Patent Publication Number: US-8970308-B2

Title: Input match network with RF bypass path

Description:
PRIORITY CLAIM 
     This application is a continuation-in-part of U.S. application Ser. No. 13/763,373 filed 8 Feb. 2013, the content of said application incorporated herein by reference in its entirety. 
    
    
     FIELD OF TECHNOLOGY 
     The present application relates to RF (radio frequency) amplifiers, in particular input match networks for RF amplifiers. 
     BACKGROUND 
     RF power amplifiers are used in a variety of applications such as base stations for wireless communication systems etc. The signals amplified by the RF power amplifiers often include signals that have a high frequency modulated carrier having frequencies in the 400 megahertz (MHz) to 4 gigahertz (GHz) range. The baseband signal that modulates the carrier is typically at a relatively lower frequency and, depending on the application, can be up to 300 MHz or higher. 
     RF power amplifiers are designed to provide linear operation without distortion. Input and output impedance matching circuits are used to match RF transistors that may have low input and output impedances (e.g., around 1 ohm or less for high power devices), to external transmission lines that provide RF signals to and from the RF transistor. These external transmission lines have characteristic impedances that are typically 50 ohms but could be any value as decided by a designer. The input and output matching circuits typically include inductive and capacitive elements that are used to provide impedance matching between the input and output of the RF power amplifier and the input and output of the RF transistor. The input and output matching circuits provide impedance matching for the signal frequencies that are amplified by the RF power amplifier, such as those in the 400 MHz to 4 GHz range. 
     The use of impedance matching circuits, however, can cause unintended consequences that occur outside of the range of signal frequencies that the impedance match is being provided for. For example, a typical output match network will include a blocking capacitor for blocking DC. The blocking capacitor in combination with the RF transistor drain bias inductance creates a low frequency resonance. This low frequency resonance causes the impedance in the low frequency region to increase. As a result, the frequency response of the RF power amplifier has a low frequency gain spike. Such a spike can appear anywhere from a few MHz to hundreds of MHz. The output of a nonlinear operation yields terms with frequencies at the sum and difference of the original signal frequencies, plus the original frequencies and multiples of the original frequencies, and those multiples are commonly referred to as harmonics. Current wireless signals have high modulation bandwidths. The second order distortion components of such wideband signals may fall in the region of the low frequency gain spike. Further, in most wireless communication applications distortion correction systems such as DPD or Digital Pre-Distortion are used. Such systems model the power amplifier, predict the non-linear performance and adjust the signal characteristics to reduce the distortion at the PA system output. The undesired high gain (or high impedance) in the baseband region due to the low frequency resonance negatively impacts the RF transistor and pre-distortion performance of the overall system. 
     A resonance in baseband frequency region causes a sharp change in gain at these low frequencies. The frequency at which the low frequency gain peak occurs is typically known as the video bandwidth of the RF power amplifier. Moreover, the magnitude of the gain peak also impacts the system performance. A higher magnitude of gain peak results in worse overall system performance. Additionally, the resonance in the baseband frequency region causes high peak voltages at the drain of RF transistors such as LDMOS (laterally-diffused metal-oxide semiconductor) transistors. These high peak voltages at the drain of the RF transistor can surpass the breakdown voltage of the device under certain conditions causing failures. Consequently, any increase of the gain peak within the low frequency baseband region can effectively reduce the ruggedness of the power device. 
     SUMMARY 
     According to an embodiment of a power circuit, the power circuit includes a RF transistor and an input match network coupled to an input to the RF transistor and to an input to the power circuit. The input match network includes a resistor, an inductor and a capacitor that are coupled together in series between the input to the RF transistor and a ground. The values of the resistor and the inductor are selected to match an input impedance of the RF transistor to a source impedance at the input to the power circuit over at least a portion of a high frequency range, wherein the value of the capacitor has a substantially negligible contribution to the match at the high frequency range. The value of the capacitor is selected so that the series combination of the resistor, the inductor and the capacitor substantially reduce the magnitude of the impedance presented to the input of the RF transistor in a low frequency range relative to the source impedance at the input to the power circuit. 
     According to another embodiment of a power circuit, the power circuit comprises an RF transistor and an input match network coupled to an input to the RF transistor and to an input to the power circuit. The input match network includes a resistor, an inductor and a first capacitor coupled together in series between the input to the RF transistor and a ground, and a second capacitor coupled in parallel with at least the resistor. The values of the resistor and the inductor are selected to match an input impedance of the RF transistor to a source impedance at the input of the power circuit over at least a portion of a high frequency range. The value of the first capacitor is selected so that the series combination of the resistor, the inductor and the first capacitor reduce the magnitude of the impedance presented to the input of the RF transistor in a low frequency range relative to the source impedance at the input of the power circuit. The value of the second capacitor is selected so that the resistor is bypassed over at least a portion of the high frequency range. 
     According to an embodiment of an RF power amplifier, the RF power amplifier comprises an input configured to receive a RF signal having a RF signal bandwidth, a LDMOS transistor configured to amplify the RF signal and an input match network coupled to the input of the RF power amplifier and a gate of the LDMOS transistor. The input match network includes a resistor, an inductor and a first capacitor coupled together in series between the input of the RF power amplifier and a ground, and a second capacitor coupled in parallel with at least the resistor. The values of the resistor and the inductor are selected to match an impedance at the gate of the LDMOS transistor to a source impedance at the input of the RF power amplifier over at least a portion of the RF signal bandwidth. The value of the first capacitor is selected so that the input match network substantially reduces the magnitude of the impedance presented to the gate of the LDMOS transistor in a baseband frequency range relative to the source impedance at the input of the RF power amplifier. The value of the second capacitor is selected so that the resistor is bypassed over at least a portion of the RF signal bandwidth. 
     According to another embodiment of a power circuit, the power circuit comprises an RF transistor and an input match network coupled to an input to the RF transistor and to an input to the power circuit. The input match network includes a resistor, an inductor and a first capacitor coupled together in series between the input to the RF transistor and a ground, and a second capacitor coupled in parallel with at least the resistor. The value of the resistor is in the milliohm range, the value of the inductor is in the pH range, the value of the first capacitor is in the nF range, and the value of the second capacitor is in the pF range. 
     Those skilled in the art will recognize additional features and advantages upon reading the following detailed description, and upon viewing the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The elements of the drawings are not necessarily to scale relative to each other. Like reference numerals designate corresponding similar parts. The features of the various illustrated embodiments can be combined unless they exclude each other. Embodiments are depicted in the drawings and are detailed in the description which follows. 
         FIG. 1  illustrates a conventional power circuit. 
         FIG. 2  illustrates an embodiment of a power circuit. 
         FIG. 3  illustrates the impedance at the input of the RF transistor as a function of frequency with the power circuits illustrated in  FIG. 1  and  FIG. 2 . 
         FIG. 4  illustrates embodiments of the gain response of a power circuit as a function of frequency and blocking capacitor value. 
         FIG. 5  illustrates embodiments of the gain response of a power circuit as a function of frequency and blocking capacitor value. 
         FIG. 6  illustrates the gain response peak attenuation as a function of frequency for the power circuits illustrated in  FIG. 1  and  FIG. 2 . 
         FIG. 7  illustrates another embodiment of a power circuit. 
         FIG. 8  illustrates yet another embodiment of a power circuit. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  illustrates a conventional power circuit  100  that includes a RF transistor  110 , an input match network  120  and an output match network  130 . Input match network  120  provides impedance matching between input terminal (IN) and a gate (G) of RF transistor  110 , and output match network  130  provides impedance matching between a drain (D) of RF transistor  110  and output terminal (OUT). The impedance matching is provided for a desired RF signal bandwidth (also referred to as the RF region). The RF region for signals that are amplified by power circuit  100  can be in the 400 MHz to 4 GHz range. The RF region for different applications can be lower or higher than the aforementioned range. 
     Input match network  120  includes a matching capacitor C IN  having one terminal coupled to ground. A branch L IN2  of the input match network  120  is coupled to the other terminal of C IN  and to the gate (G) of an RF transistor  110 . A branch L IN1  couples the input terminal (IN) of the power circuit  100  to the terminal of L IN2  and the other terminal of C IN . The branches L IN1  and L IN2  of the input match network  120  are typically implemented as bond wires, ribbons, etc. The input match network  120  uses selected values for C IN  and L IN2  to match an input impedance of RF transistor  110  to the terminal (IN) impedance. In this illustration in  FIG. 1 , C IN =10 to 100 pF, L IN1 =100 to 200 pH and L IN2 =100 to 200 pH. RF transistor  110  is a LDMOS (laterally-diffused metal-oxide semiconductor) transistor rated at 100 Watts. The magnitude of the impedance presented to the gate (G) of RF transistor  110  is designated by Z IN1  and the reference arrow illustrates that this impedance is provided to the gate (G) terminal of the RF transistor  110 . The impedance Z IN1  will be discussed in reference to  FIG. 3  and  FIG. 6 . 
     The output match network  130  includes a blocking capacitor C OUT  and a branch L OUT1 . L OUT1  is coupled to the drain (D) of the RF transistor  110  and to one terminal of C OUT . The other terminal of C OUT  is coupled to ground. A branch L OUT2  couples the drain (D) of RF transistor  110  and one terminal of L OUT1  to the terminal (OUT) of the power circuit  100 . The source (S) of the RF transistor  110  is coupled to ground. The branches L OUT1  and L OUT2  of the output match network  130  can be implemented in various different ways such as bond wires, ribbons, etc. The output match network  130  uses C OUT  and L OUT1  to match an output impedance at the drain (D) of RF transistor  110  to a terminal (OUT) impedance within the RF region. 
     The output match network  130  provides high frequency impedance matching over the RF region but may also result in an undesirable low frequency gain peak outside of the RF region which corresponds to the low frequency resonance. The blocking capacitor C OUT  cuts off DC, and the combination of LC components (e.g., including, but not limited to, inductance from voltage connections external to power circuit  100 , branch L OUT1  and branch L OUT2 ) with the DC blocking capacitance form resonances, resulting in a high gain peak and high peak voltages at the drain of RF transistor  110  at frequencies in the low frequency baseband region. 
     This high gain peak in the baseband frequency region can cause the peak drain voltage of RF transistor  110  to surpass the breakdown voltage of the device under certain conditions, e.g., when unintended system artifacts appearing in the region of the low frequency gain peak are strongly amplified, when a baseband component of a broadband signal that is coincident with the low frequency gain response peak is strongly amplified by the gain peak, etc. It will be appreciated that this problem cannot be completely solved with a more complex matching circuit that suppresses low frequency signals before they reach the output terminal, since the excessively high low-frequency signals will still be present at the output of the transistor. 
       FIG. 2  illustrates an embodiment of a power circuit  200 . The power circuit  200  includes a RF transistor  210 , an input match network  220  and an output match network  230 . Input match network  220  provides impedance matching between input terminal (IN) and a gate (G) of RF transistor  210 , and output match network  230  provides impedance matching between a drain (D) of RF transistor  210  and output terminal (OUT). The impedance matching is provided for a desired RF signal bandwidth or RF region. In other embodiments, the impedance matching is provided for at least a portion of a desired RF signal bandwidth or RF region. The RF region for signals that are amplified by power circuit  200  can be in the 400 MHz to 4 GHz range. The RF region for different applications can be lower or higher than the aforementioned range. 
     The input match network  220  is coupled between an input inductor L IN1  of power circuit  200  and a gate (G) of RF transistor  210 . In various embodiments, the input inductor L IN1  is implemented as bond wires, ribbons etc. which couple the input match network  220  to a terminal (IN) of the power circuit  200 . In many cases L IN1  is a part of the input match network. The output match network  230  is coupled between a drain (D) of RF transistor  210  and an output inductor L OUT2  of the power circuit  200 . In various embodiments, the output inductor L OUT2  is implemented as bond wires, ribbons etc. which couple the output match network  230  to a terminal (OUT) of the power circuit  200 . In many cases L OUT2  is a part of the output match network. Output match network  230  is similar to output match network  130  discussed in reference to  FIG. 1 . The magnitude of the impedance presented to the gate (G) of RF transistor  210  is designated by Z IN2  and the reference arrow illustrates that this impedance is provided to gate (G) of RF transistor  210 . The impedance Z IN2  will be discussed in reference to  FIG. 3  and  FIG. 6 . 
     In various embodiments, RF transistor  210  can be a power transistor such as a MOSFET (metal-oxide semiconductor field-effect transistor), DMOS (double-diffused metal-oxide semiconductor) transistor, GaN HEMT (gallium nitride high electron mobility transistor), GaN MESFET (gallium nitride metal-semiconductor field-effect transistor), LDMOS transistor, etc. and more generally any type of RF transistor device. RF transistor  210  and the complete power circuit  200  can be a multi-carrier amplifier, a multiband amplifier, an LTE (long term evolution) compliant amplifier, a WCDMA (wideband code division multiple access) compliant amplifier, an 802.11(x) compliant amplifier, etc. 
     Input match network  220  includes a blocking capacitor C IN , a resistance R IN  and an inductance L IN2 . In this embodiment, C IN , R IN  and L IN2  are coupled in series between a gate (G) of RF transistor  210  and ground. Although the illustrated embodiment shows this series connection with one terminal of C IN  coupled to ground and one terminal of L IN2  coupled to a gate (G) of RF transistor  210  with R IN  coupled between a second terminal of L IN2  and a second terminal of C IN , in other embodiments, C IN , R IN  and L IN2  can be coupled between ground and gate (G) of RF transistor  210  in other suitable configurations. 
     The branch L IN2  of input match network  220  can be implemented as bond wires, ribbons, etc. In various embodiments, the branch L IN2  can be implemented as other suitable inductors. The blocking capacitor C IN  of the input match network  220  can be implemented as a discrete component separate from RF transistor  210  or can be integrated with RF transistor  210  on the same die. Resistance R IN  and inductance L IN2  can be implemented as discrete components or distributed components separate from the RF transistor  210  or can be integrated with the RF transistor  210  on the same die. In one embodiment, resistance R IN  and inductance L IN2  provide input match compensation for the parasitic capacitances of RF transistor  210  including, but not limited to, the gate (G) to source (S) capacitance of RF transistor  210 . The input match network  220  can have other configurations which are within the scope of the embodiments described herein. 
     Input match network  220  provides a low impedance at the gate (G) of RF transistor  210  over the baseband frequency range, e.g., between 0 to 300 MHz, and reduces a gain response peak within this frequency range, which may result in lower peak voltages at the drain (D) of RF transistor  210 . Input match network  220  provides an impedance match between terminal (IN) and a gate (G) of RF transistor  210  over a range of signal frequencies that are amplified by power circuit  200 . In other embodiments, input match network  220  provides an impedance match between terminal (IN) and a gate (G) of RF transistor  210  over at least a portion of a range of signal frequencies that are amplified by power circuit  200 . In one embodiment, the range of signal frequencies that are amplified by power circuit  200  are in the 1 to 3 GHz range. In another embodiment, the range of signal frequencies that are amplified by power circuit  200  are in the 400 MHz to 4 GHz range. 
     The values of resistor R IN  and inductor L IN2  are selected to create a match between an input impedance of RF transistor  210  and impedance at the terminal (IN) of power circuit  200 . Within the range of signal frequencies that are in the 400 MHz to 4 GHz range, the value of capacitor C IN  has a substantially negligible contribution to the impedance match. The value of capacitor C IN  is selected so that the series combination of resistor R IN , inductor L IN2  and capacitor C IN  substantially reduces the magnitude of the impedance presented to the gate (G) of RF transistor  210  over the baseband frequency range. In other embodiments, for different applications, the baseband frequency range can include frequencies that are greater than 300 MHz, and the RF signal bandwidth or RF region for signals that are amplified by power circuit  200  can include frequencies that are lower than 400 MHz or frequencies that are greater than 4 GHz. In the embodiment illustrated in  FIG. 2 , C IN =2 nF, R IN =0.3 ohms and L IN 2=70 pH. Because the capacitance value of C IN  used in input match network  220  may depend on the application in which the power circuit  200  is used, in other embodiments, capacitor C IN  can have other suitable values. 
       FIG. 3  illustrates a baseband impedance presented to gate (G) of RF transistor  110 / 210  as a function of frequency and respectively with input match networks  120 / 220 . Curve  310  illustrates the impedance at gate (G) of RF transistor  110  for power circuit  100  (refer to Z IN1  in  FIG. 1 ) with conventional input match network  120 . Curve  320  illustrates the impedance at gate (G) of RF transistor  210  for power circuit  200  (refer to Z IN2  in  FIG. 2 ) for an embodiment of input match network  220  where C IN  has a value of 2 nF. The terminals (IN) of power circuits  100 / 200  have a characteristic impedance of 50 ohms. 
     Over the baseband frequency range of 0 to 300 MHz, input match network  220  provides a much lower input impedance to gate (G) of RF transistor  210  than conventional input match network  120  provides to gate (G) of RF transistor  110 . Curve  310  illustrates that with input match network  120 , a maximum impedance is presented at gate (G) of RF transistor  110  of about 48 ohms (at 1 MHz). With input match network  220 , curve  320  illustrates that a maximum impedance is presented at gate (G) of RF transistor  210  of about 6 ohms (at 1 MHz). 
     Overall the maximum impedance presented to gate (G) of RF transistor  110 / 210 , over the low-frequency range, in comparison to the characteristic input impedance at terminal (IN) of power circuits  100 / 200  is much lower with input match network  220  than with input match network  120 . In the illustrated embodiment, for a 100 W RF power transistor, a ratio of the magnitude of impedance Z IN2  presented to gate (G) of RF transistor  210  to the source impedance of 50 ohms for power circuit  200  ranges from approximately 0.02 to 0.12 over a 1 to 300 MHz frequency range. In other embodiments, the ratio of the magnitude of impedance Z IN2  presented to gate (G) of RF transistor  210  to the source impedance of 50 ohms for power circuit  200  has a maximum value of 0.4 over the 1 to 300 MHz frequency range. In other embodiments, a ratio of the magnitude of impedance Z IN2  presented to gate (G) of RF transistor  210  to the source impedance of 50 ohms for power circuit  200  ranges from approximately 0.02 to 0.4 over a 1 to 300 MHz frequency range. 
     The input match network  220  provides an impedance match between terminal (IN) and gate (G) of RF transistor  210  at the intended RF operating frequency, which is approximately 2 GHz in this illustration, and the value of capacitor C IN  has a substantially negligible contribution to the impedance match. As  FIG. 3  illustrates, beginning at 1 GHz, the impedances presented to gate (G) of RF transistors  110 / 210  by input match networks  120 / 220  are approximately equivalent. The value of capacitor C IN  is selected so that the series combination of resistor R IN , inductor L IN2  and capacitor C IN  substantially reduce the magnitude of the impedance presented to the input of RF transistor  210  over the baseband frequency range at the terminal (IN) of power circuit  200 . Curve  320  illustrates that with input match network  220 , a maximum impedance is presented to gate (G) of RF transistor  210  of about 6 ohms (at 1 MHz) which is significantly lower than the source impedance of 50 ohms at the terminal (IN) of power circuit  200 . In other embodiments, a maximum impedance presented to gate (G) of RF transistor  210  is equal to or less than 20 ohms in the low frequency region. This is illustrated in  FIG. 3 . In other embodiments, the maximum impedance presented to gate (G) of RF transistor  210  is equal to or less than 20 ohms for frequencies ranging from 1 MHz up to at least one-third of the intended RF operating frequency. Note that curve  310  shows an impedance magnitude that is close to 50 ohms over the low-frequency range. This indicates that the conventional matching circuit, which is a low pass circuit, is having little or no effect on signals in the low-frequency region. For a larger device, for instance a 200 W RF transistor, the maximum impedance presented to the gate of the transistor in the low frequency region may be approximately 10 ohms. Similarly, for a 50 W RF transistor, the maximum impedance presented to the gate of the transistor in the low frequency region may be approximately 40 ohms. 
       FIGS. 4 and 5  illustrate the gain response (dB) of power circuit  200  as a function of the value of blocking capacitor C IN  and frequency.  FIG. 4  illustrates a frequency range from 0 to 700 MHz which includes the low frequency baseband range of 1 to 300 MHz.  FIG. 5  illustrates a frequency range from 0 to 3 GHz which includes both the baseband frequency range and the RF region of operation. Referring to  FIG. 4 , curve  410  represents a blocking capacitor C IN  value of 100 pF, curve  420  represents a blocking capacitor C IN  value of 500 pF, curve  430  represents a blocking capacitor C IN  value of 1.2 nF, and curve  440  represents a blocking capacitor C IN  value of 2 nF. Referring to  FIG. 5 , curve  510  represents a blocking capacitor C IN  value of 100 pF, curve  520  represents a blocking capacitor C IN  value of 500 pF, curve  530  represents a blocking capacitor C IN  value of 1.2 nF, and curve  540  represents a blocking capacitor C IN  value of 2 nF. Referring to  FIG. 4  and  FIG. 5 , for capacitor C IN  values of 100 pF (curve  410 / 510 ) and 2 nF (curve  440 / 540 ), for increasing values of C IN  the gain and corresponding gain peak in the baseband range is significantly reduced while the impact of the value of C IN  on the gain at the intended RF operating frequency, which is approximately 2 GHz, is much lower and becomes increasingly negligible at higher frequencies. 
     For a capacitor C IN  value of 2 nF (curve  440 / 540 ), the gain response peak in the low frequency baseband range of 0 to 300 MHz (see  FIG. 4 ) is about −12 dB while the gain response at the approximate intended RF operating frequency, which is approximately 2 GHz, is about 23 dB. The difference between the gain response peak in the low frequency baseband range and the gain response at the approximate intended RF operating frequency, which for this embodiment is approximately 2 GHz, is about 35 dB for a capacitor C IN  value of 2 nF. For a capacitor C IN  value of 1.2 nF (curve  430 / 530 ), the gain response peak in the low frequency baseband range of 0 to 300 MHz (see  FIG. 4 ) is about −12 dB while the gain response at the approximate intended RF operating frequency, which is approximately 2 GHz, is about 23 dB. The difference between the gain response peak in the low frequency baseband range and the gain response in the approximate center of the high frequency RF region for a capacitor C IN  value of 1.2 nF is about 35 dB. For capacitor C IN  values of 2.0 nF and 1.2 nF, the gain response peak reduction in the low frequency baseband range has resulted in a difference between the gain response peak in the low frequency baseband range and the gain response at the approximate intended RF operating frequency of about 35 dB. In another embodiment, the difference between the gain response peak in the low frequency baseband range and the gain response at the intended RF operating frequency is about 28 dB. In other embodiments, the high frequency RF region and the intended RF operating frequency can be in the 400 MHz to 4 GHz range. The RF region and the intended RF operating frequency for different applications can be lower or higher than the aforementioned range. Furthermore, the low frequency baseband range for different applications can extend to frequencies that are greater than 300 MHz. 
       FIG. 6  illustrates measured results of the gain response of conventional power circuit  100  and an embodiment of power circuit  200  in the baseband frequency range of 0 to 400 MHz. Curve  610  illustrates the gain response for power circuit  100  and curve  620  illustrates the gain response for power circuit  200 . In this embodiment of power circuit  200 , input match circuit  220  has a blocking capacitor C IN  value of 2 nF. For conventional power circuit  100 , curve  610  illustrates a gain peak of −2 dB at 217 MHz (refer to m1). For the illustrated embodiment of power circuit  200 , curve  620  illustrates a gain peak of −15 dB at 235 MHz (refer to m2). Compared to power circuit  100 , power circuit  200  reduces the gain in the baseband region from −2 dB to −15 dB and increases the frequency at which the gain peak occurs from 217 MHz to 235 MHz. Such a reduction in gain peak results in improved DPD system performance for power circuit  200  and improved ruggedness for RF transistor  210 . 
       FIG. 7  illustrates another embodiment of a power circuit  300 . The power circuit  300  shown in  FIG. 7  is similar to the power circuit  200  shown in  FIG. 2 . Different than the power circuit  200  shown in  FIG. 2 , the input match network  220  of the power circuit  300  shown in  FIG. 7  further includes a bypass capacitor C bypass  connected in parallel with the blocking capacitor C IN  and the resistor R IN  of the input match network  220 . The input match network  220  enables broader RF bandwidth for the power circuit  300  and tailors the low frequency (baseband) response of the RF transistor  210  as previously explained herein. However, the resistor R IN  of the input match network  220  can cause slight matching losses over the RF region for signals that are amplified by the power circuit  300 . Such matching losses reduce the RF gain of the power circuit  300  if unmitigated. 
     The bypass capacitor C bypass  connected in parallel with the blocking capacitor C IN  and the resistor R IN  of the input match network  220  provides an RF bypass path around the resistor R IN  over at least a portion of the high frequency (RF) region for signals that are amplified by power circuit  300 , effectively removing the resistor R IN  over at least a portion of the RF region of interest. According to this embodiment, the resistor R IN  of the input match network  220  has no adverse effect on the gain of the power circuit  300  over at least a portion of the RF signal region but is available (i.e. not bypassed) over the low frequency baseband region to advantageously tailor the baseband characteristics of the power circuit  300  as previously described herein. 
       FIG. 8  illustrates yet another embodiment of a power circuit  400 . The power circuit  400  shown in  FIG. 8  is similar to the power circuit  300  shown in  FIG. 7 . Different than the power circuit  300  shown in  FIG. 7 , the bypass capacitor C bypass  is connected in parallel with only the resistor R IN  of the input match network  220 . This way, the physical size of the bypass capacitor C bypass  can be reduced since it is not directly connected to ground. 
     In either case, the blocking capacitor C IN  of the input match network  220  is a high value capacitor that provides a low impedance path to the RF signal envelope. In one embodiment, the blocking capacitor C IN  is in the nF range e.g. multiples of nF in value. The bypass capacitor C bypass  of the input match network  220  is a bypass capacitor that bypasses the resistor R IN  of the input match network  220  over at least a portion of the RF region for signals that are amplified and reduces matching losses caused by the resistor R IN . In one embodiment, the bypass capacitor C bypass  is in the pF range e.g. multiples of pF in value. The resistor R IN  of the input match network  220  can be in the milliohm range e.g. multiples of milliohms in value, and the inductor L IN2  of the input match network  220  can be in the pH range e.g. multiples of pH in value. The components L IN2 , R IN , C IN , and C bypass  of the input match network  220  can be integrated on a single semiconductor IC such as a silicon IC or other type of semiconductor IC. 
     Spatially relative terms such as “under”, “below”, “lower”, “over”, “upper” and the like, are used for ease of description to explain the positioning of one element relative to a second element. These terms are intended to encompass different orientations of the device in addition to different orientations than those depicted in the figures. Further, terms such as “first”, “second”, and the like, are also used to describe various elements, regions, sections, etc. and are also not intended to be limiting. Like terms refer to like elements throughout the description. 
     As used herein, the terms “having”, “containing”, “including”, “comprising” and the like are open ended terms that indicate the presence of stated elements or features, but do not preclude additional elements or features. The articles “a”, “an” and “the” are intended to include the plural as well as the singular, unless the context clearly indicates otherwise. 
     With the above range of variations and applications in mind, it should be understood that the present invention is not limited by the foregoing description, nor is it limited by the accompanying drawings. Instead, the present invention is limited only by the following claims and their legal equivalents.