Patent Publication Number: US-8995565-B2

Title: Distortion control device and method

Description:
TECHNICAL FIELD 
     The present invention relates to a distortion control device and method. 
     BACKGROUND ART 
     The transmission-side circuit arrangement of a mobile terminal apparatus based on a general W-CDMA scheme will be described with reference to  FIG. 9 . A baseband signal comprises an in-phase component (to be referred to as an I signal) and a quadrature component (to be referred to as a Q signal) in quadrature modulation. A digital baseband unit  112  generates this signal. RRC (Raised Root Cosine) roll-off filters  110  and  111  for waveform shaping band-limit the I and Q signals. The processing so far is digital signal processing. 
     D/A converters  108  and  109  then respectively convert the I and Q signals into analog signals. A known quadrature modulator  106  performs quadrature modulation for a local signal with these analog signals. The high-frequency signal generated as a result of this operation is input to a variable gain amplifier (VGA)  105 , which then amplifies it to a predetermined level in accordance with the gain control signal output from the digital baseband unit  112  or the analog signal converted from the gain control signal by a D/A converter  107 . 
     The high-frequency signal amplified by the variable gain amplifier  105  contains many spurious components. A bandpass filter (BPF)  104  removes these spurious components. The resultant high-frequency signal is then amplified by a power amplifier (PA)  102  and transmitted from an antenna  101 . Although a power supply  103  drives the power amplifier  102 ,  FIG. 9  shows the voltage of the power supply  103  as a fixed voltage. 
       FIG. 10  shows the arrangement of a circuit for generating a baseband signal in a currently commercialized W-CDMA (referred to as R99: Release 99) mobile terminal apparatus. Reference symbol DPCCH denotes a control channel, which is a binary signal of ±1. A multiplier  133  multiplies this signal by a spreading code Cc (which is also a binary signal of ±1). A multiplier  134  then multiplies the signal by a weighting factor βc. On the other hand, reference symbol DPDCH denotes a data channel, which is a binary signal of ±1 as in the case with DPCCH. A multiplier  130  multiplies this signal by a spreading code Cd (which is also a binary signal of ±1). A multiplier  131  then multiplies the resultant signal by a weighting factor βd. 
     In the R99 system, a baseband signal comprises only these two-system signals in reality. A scrambler  138  multiplies this signal by a scramble code, and then outputs real and imaginary parts as I and Q signals, respectively. Reference numerals  132  and  135  denote combiners;  136 , a multiplier which multiplies j representing an imaginary number; and  137 , an adder which adds a real part and an imaginary part. 
       FIG. 11A  shows the constellation of baseband signals (loci on an IQ plane) after they pass through the RRC roll-off filters  110  and  111 . Referring to  FIG. 11A , the ratios between the values of weighting factors β are βc=8/15 and βd=15/15. In the constellation chart, the white dotted line circle is a circle whose radius is defined by the RMS (root mean square) value of signal amplitudes, and the black solid line circle is a circle whose radius is defined by a peak value. According to R99, since the number of code channels constituting a baseband is only two, i.e., DPCCH and DPDCH, the ratio (PAR: Peak Average Ratio) between the peak value and the RMS value is small. When this value is represented by dB, the resultant value is about 3.3 dB at most. 
       FIG. 12  shows the arrangement of a circuit which generates a baseband signal based on an HSDPA (High Speed Downlink Packet Access) (R5: Release 5) scheme which is expected to be commercialized in the near future. The same reference numerals as in  FIG. 10  denote the same parts in  FIG. 12 . In Release 5, HS(High Speed)-DPCCH, which is a new control channel, is additionally provided as a response channel for a high speed downlink data channel, as shown in  FIG. 12 . An HS-DPCCH signal is an uplink control channel for HSDPA, and is a binary signal of ±1. A multiplier  139  multiplies this signal by a spreading code Chs (which is also a binary signal of ±1). A multiplier  140  multiplies the resultant signal by a weighting factor βhs. With the addition of this HS-DPCCH, the PAR increases to about 5 dB. 
     According to HSUPA (High Speed Uplink Packet Access) (R6: Release 6) which is expected to be adopted in the future, the number of code channels greatly increases, as shown in  FIG. 13 . The same reference numerals as in  FIG. 12  denote the same parts in  FIG. 13 . In addition to DPDCH, high-speed data channels E-DPDCH 1  to E-DPDCH 4  are additionally provided, which are respectively spread by unique spreading codes Ced, 1  to Ced, 4  (multipliers  141 ,  143 ,  145 , and  147 ) and respectively weighted by unique weighting factors βed, 1  to βed, 4  (multipliers  142 ,  144 ,  146 , and  148 ). 
     In addition, a control channel E-DPCCH for controlling these communications is additionally provided. The control channel E-DPCCH is spread by a unique spreading code Cec (a multiplier  149 ) and weighted by a unique weighting factor βec (a multiplier  150 ).  FIG. 11B  shows a constellation in a case in which the ratios between weighting factors are βc=βhs=βec=8/15, βd=0, βed 1 =βed 2 =15/15, and βed 3 =βed 4 =11/15. Compared with R99, the gap between the peak value and the RMS value is large, and the PAR is about 7 dB. That is, the PAR is larger than that in R99 by as large as 4.6 dB. 
     Compared with R99, therefore, HSUPA cannot meet the adjacent channel leakage power standard when amplification is performed by the same amplifier, because a large distortion occurs at an amplitude peak, unless the transmission power is decreased. A dB value indicating how much the transmission power should be decreased to meet the adjacent channel leakage power standard is called a back-off. Since R99 is currently in practical use, a dB value indicating how much the transmission power is decreased as compared with R99 to obtain the same adjacent channel leakage power as that in R99 is called a back-off relative to R99. This value will be simply referred to as a back-off hereinafter. 
     DISCLOSURE OF INVENTION 
     Problem to be Solved by the Invention 
     A PAR value is an index which is analogous to a back-off, but does not always coincide with it. This is because, the back-off changes depending on the probability distribution of peak values. A back-off value is almost determined by a combination of β (weighting factors). In R99 and HSDP, the number of combinations of β is not very large. In HSUPA, however, there are several million combinations of β because of a great increase in the number of code channels. It is impossible to generate a table by calculating back-off values for all the combinations. 
     The technique disclosed in reference 1 (Japanese Patent Laid-Open No. 2005-252388) is a technique conforming to HSDPA in 3GPP (3rd Generation Partnership Project). As shown in  FIG. 12 , this technique considers only one to three channels, and is designed to simply reduce the maximum transmission power in a plurality of steps on the basis of a gain factor ratio. The gain factors in this case are βd, βc, and βhs in  FIG. 12 . This technique is designed to decrease transmission power by determining a plurality of reduction amounts in accordance with these combinations, thereby improving an ACLR (Adjacent Channel Leakage Power Ratio). This technique is also a scheme using a table in which reduction amounts are determined in correspondence with the above combinations of gain factors. 
     In the case of HSUPA, as described above, since the number of code channels greatly increases, several million combinations are required, and it is impossible to generate a table. The same applies to the technique disclosed in reference 2 (Japanese Patent Laid-Open No. 2005-318266). When the number of code channels greatly increases as in HSUPA, this technique is inadequate. 
     The present invention has been made to solve this problem, and has as its object to provide a distortion control device and method which can easily control transmission power to improve an ACLR without using any table. 
     Means of Solution to the Problem 
     A distortion control device according to the present invention comprises waveform analyzing means for calculating an estimated value of a back-off value required by a power amplifier, which amplifies a high-frequency signal generated from a baseband signal to a predetermined transmission power, by analyzing a waveform of the baseband signal, and control means for controlling at least one of an amplitude of high-frequency power input to the power amplifier and supply power of the power amplifier on the basis of an estimated value calculated by the waveform analyzing means. 
     A distortion control method according to the present invention comprises the steps of calculating an estimated value of a back-off value required by a power amplifier, which amplifies a high-frequency signal generated from a baseband signal to a predetermined transmission power, by analyzing a waveform of the baseband signal, and controlling at least one of an amplitude of high-frequency power input to the power amplifier and supply power of the power amplifier on the basis of a calculated estimated value. 
     Effects of the Invention 
     According to the present invention, since the estimated value of a back-off value is calculated by analyzing the waveform of a baseband signal, there is no need to generate a table in advance by calculating a back-off value in advance for each combination of code channels. The present invention can therefore be applied to even a case in which the number of code channels greatly increases, and can effectively prevent an increase in adjacent channel leakage power due to a signal obtained by multiplexing these code channels. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a block diagram showing the arrangement of an exemplary embodiment of the present invention; 
         FIG. 2  is a block diagram showing an example of the arrangement of a waveform analyzing unit in  FIG. 1 ; 
         FIG. 3  is a graph for explaining the operation of a maximum power reducer in  FIG. 1 ; 
         FIG. 4  is a block diagram showing the arrangement of another exemplary embodiment of the present invention; 
         FIG. 5  is a block diagram showing the arrangement of the main part of a digital baseband unit in  FIG. 1 ; 
         FIG. 6  is a block diagram showing an example of the arrangement of a transmitter which performs power supply control on a power amplifier; 
         FIG. 7  is a block diagram showing the arrangement of another exemplary embodiment of the present invention when the present invention is applied to the transmitter shown in  FIG. 6 ; 
         FIG. 8  is a graph for explaining the effects of the exemplary embodiment shown in  FIG. 7 ; 
         FIG. 9  is a block diagram showing the arrangement of the transmission-side circuit of a mobile terminal apparatus based on a general W-CDMA scheme; 
         FIG. 10  is a block diagram of a circuit which generates a baseband signal in the mobile terminal apparatus based on the W-CDMA (R99) scheme; 
         FIG. 11A  is a view showing a constellation which is the loci of baseband signals on an IQ plane in the W-CDMA (R99) scheme; 
         FIG. 11B  is a view showing a constellation which is the loci of baseband signals on an IQ plane in the HSUPA (R6) scheme; 
         FIG. 12  is a block diagram showing a circuit which generates a baseband signal in a mobile terminal apparatus based on the HSDPA (R5) scheme; and 
         FIG. 13  is a block diagram showing a circuit which generates a baseband signal in a mobile terminal apparatus based on the HSUPA (R6) scheme. 
     
    
    
     BEST MODE FOR CARRYING OUT THE INVENTION 
     The principle of the exemplary embodiments of the present invention will be described first. The exemplary embodiment of the present invention uses a technique called Cubic Metric (to be simply referred to as CM) as a method using no table like that described above. The following is a calculation method based on this CM method. 
     A transmission waveform x(t) is given by
 
 x ( t )= V ( t )·cos {ω c·t +φ( t )}
 
where V(t) is an amplitude, and φ(t) is a phase. In CM, only an amplitude is used.
 
     First of all, RCM (Raw Cubic Metric) is defined.
 
RCM=20·log 10  {rms[ V   3   norm ( t )]}
 
where V norm (t) is given by
 
 V   norm ( t )=| V ( t )|/rms[ V ( t )]
 
As a consequence, RCM is given by
 
     
       
         
           
             
               
                 
                   
                     
                       
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     The numerator in the log is the sextic moment of an amplitude probability density function, and the denominator is the cube of average power (quadratic moment). Therefore, RCM is a variable determined when at least an amplitude probability density function is determined. With this setting, CM is obtained as follows:
 
CM={RCM(Target)−RCM( R 99)}/κ
 
     In this case, RCM(Target) is the RCM of a baseband arrangement for which CM is to be calculated, and RCM(R99) is the RCM of the R99 system. The value of RCM(R99) is almost 1.52. The value of κ is experimentally determined to an optimal value. For example, κ=1.56 and κ=1.88 may be switched depending on a baseband arrangement. The CM value calculated in this manner indicates an excellent approximation as a back-off value relative to R99, and hence is used to perform back-off control and reduce distortion in the exemplary embodiment of the present invention. 
     The exemplary embodiment of the present invention which uses the above CM method will be described below specifically with reference to the accompanying drawings.  FIG. 1  shows the arrangement of the transmitter of a mobile terminal apparatus based on the W-CDMA scheme according to an exemplary embodiment of the present invention. This transmitter comprises an antenna  1 , a power amplifier (PA)  2 , a power supply  3 , a bandpass filter (BPF)  4 , a variable gain amplifier  5 , a quadrature modulator (frequency converter)  6 , D/A converters  7 ,  8 , and  9 , RRC roll-off filters  10  and  11 , a digital baseband unit  12 , a waveform analyzing unit  13 , and a maximum power reducer (a maximum power reducer and gain control means)  14 . The waveform analyzing unit  13  and the maximum power reducer  14  constitute a distortion control device  60  as a characteristic feature of this exemplary embodiment. 
     A baseband signal comprises an in-phase component (to be referred to as an I signal) and a quadrature component (to be referred to as a Q signal) in quadrature modulation. The digital baseband unit  12  generates this signal. The RRC roll-off filters  10  and  11  for waveform shaping band-limit the I and Q signals. The processing so far is digital signal processing. 
     The D/A converters  8  and  9  then respectively convert the I and Q signals into analog signals. The known quadrature modulator  6  performs quadrature modulation for a local signal with these analog signals. The high-frequency signal generated as a result of this operation is input to the variable gain amplifier  5 , which then amplifies it to a predetermined level in accordance with the gain control signal supplied from the digital baseband unit  12  to the variable gain amplifier  5  via the distortion control device  60  and the D/A converter  7 . 
     The high-frequency signal amplified by the variable gain amplifier  5  contains many spurious components. The bandpass filter  4  removes these spurious components. The resultant high-frequency signal is then amplified to a predetermined transmission power by the power amplifier  2  and transmitted from the antenna  1 . In practice, circuits such as an isolator, a duplexer, and an antenna switch are arranged between the power amplifier  2  and the antenna  1 . These components are not directly associated with this exemplary embodiment, and hence are not illustrated in  FIG. 1 . Although the power supply  3  drives the power amplifier  2 ,  FIG. 1  shows the voltage of the power supply  3  as a fixed voltage. 
     The distortion control device  60  will be further described. As described above, the distortion control device  60  comprises the waveform analyzing unit  13  and the maximum power reducer  14 . The waveform analyzing unit  13  receives the I and Q baseband signals output from the RRC roll-off filters  10  and  11 , performs waveform analysis, and calculates and outputs the estimated value of a required back-off relative to R99 as a result of the waveform analysis. In this case, as a calculation method, the CM method is used. A concrete implementation method for this method will be described with reference to  FIG. 2 . 
     Square circuits (Square)  20  and  21  respectively square the I and Q signals input from the left side in  FIG. 2 . An adder (Add)  22  adds these two signals (I 2 , Q 2 ) and obtains V 2 (I 2 +Q 2 ) as the square of the amplitude. A mean circuit (Mean)  23  calculates, for example, the mean of W-CDMA data corresponding to one slot. A cube circuit (Cube)  25  cubes the resultant value. This result is E[V 2 ] 3 . This value is the cube of the quadrature moment of the amplitude probability density function. In this case, E[x] represents an expected value of x. 
     A cube circuit  24  cubes V 2  in advance. A mean circuit  26  calculates, for example, the mean of W-CDMA data corresponding to one slot. This result is E[V 6 ]. This value is the sextic moment of an amplitude probability density function. A divider (C=A/B)  27  divides E[V 6 ] described above by E[V 2 ] 3 . An obtained value C is given by
 
 C=E[V   6   ]/E[V   2 ] 3  
 
     An estimated value calculation unit  28  obtains the dB value of power of the value of C, subtracts a value (offset value) ref as in the case of R99 from the dB value, and divides the value by a predetermined constant κ, thereby outputting the resultant value as the estimated value (dB) of the back-off. In practice, the value obtained by increasing this value in increments of 0.5 dB and subtracting 1 from the resultant value is used as an MPR (Maximum Power Reduction) value. If a negative value is obtained by subtraction of 1, the dB value is set to 0. Note that ref corresponds to RCM(R99)≈1.52. 
     The maximum power reducer  14  receives the back-off value or MPR value output from the waveform analyzing unit  13 , and outputs, as an actual gain control signal, the result which limits the value of the gain control signal output from the digital baseband unit  12  so as to prevent it from exceeding the value obtained by subtracting the MPR value from the maximum value, as shown in  FIG. 3 . Referring to  FIG. 3 , reference symbol Al max  denotes the maximum value (maximum gain) of a gain control signal input to the waveform analyzing unit  13 ; A 2   max , the maximum value (reduced maximum gain) of the gain control signal output from the waveform analyzing unit  13 ; and R, the maximum width (gain reduction maximum width) of reduction by the waveform analyzing unit  13 . Note that it suffices to use another method of simply reducing a gain control signal by an MPR value. 
     With this operation, the gain of the variable gain amplifier  5  is limited to equal to or less than a value smaller than the maximum value by the MPR value. With this function, the output of the power amplifier  2  is limited to a value smaller than the maximum output by the MPR value. This can prevent an increase in adjacent channel leakage power due to a signal obtained by multiplexing many code channels, which increase is caused by the distortion of transmission power by the power amplifier. 
     An exemplary embodiment of the present invention will be described next.  FIG. 4  shows the arrangement of the transmitter of a mobile terminal apparatus based on the W-CDMA scheme according to another exemplary embodiment of the present invention. The same reference numerals as in  FIG. 1  denote the same parts in  FIG. 4 . As shown in  FIG. 5 , the transmitter of this exemplary embodiment comprises, inside a digital baseband unit  12   a , a baseband signal generating unit (level control means)  18  having a function of controlling the amplitudes of I and Q signals instead of the maximum power reducer  14 . In this exemplary embodiment, a waveform analyzing unit  13  and the baseband signal generating unit  18  constitute distortion control device. 
     The baseband signal generating unit  18  receives the MPR value output from the waveform analyzing unit  13 . The baseband signal generating unit  18  outputs the I and Q signals upon attenuating them from the planned level at which they are to be output by the following level.
 
attenuation amount=MAX{planned output level−(maximum level−MPR),0)}dB
 
where MAX{A, B} is a function which outputs larger one of A and B. In addition, an attenuation amount can be simply set as attenuation amount=MPR.
 
     With this operation, the same effect as that of the exemplary embodiment shown in  FIG. 1  can be obtained. A merit of this exemplary embodiment is that it is possible to cope with distortion of transmission power due to the variable gain amplifier  5  as well as a power amplifier  2 . Note that the maximum power reducer  14  in  FIG. 1  and the baseband signal generating unit  18  in  FIG. 5  are common in that control is performed to attenuate the high-frequency power input to the power amplifier  2 . 
     In the above two exemplary embodiments, adjacent channel leakage power due to distortion is prevented by calculating a back-off value or MPR value by analyzing the waveform of a baseband, and reducing the transmission power by a value corresponding to the transmission power. This will always reduce transmission power. In this case, the radius of a cell in which a base station provides services decreases. 
     If, for example, the transmission power decreases by 1 dB, the radius of the cell becomes 0.89 times based on the assumption of free-space propagation. In terms of area, the cell becomes 0.79 times, i.e., decreases by about 20%. Therefore, a reduction in maximum transmission power by 1 dB means a reduction in the area of the cell by 20%. In other words, it is necessary to install base stations by 20% more. The operator requires extra expenditure, which is reflected in the call charge for the user in the end. It is therefore desirable to perform distortion control without decreasing transmission power if possible. 
       FIG. 6  shows an example of the arrangement of a transmitter which performs power supply control on a power amplifier. The same reference numerals as in  FIG. 1  denote the same parts in  FIG. 6 . Assume that a power supply to be used is a variable voltage power supply  3   a . A digital baseband unit  12   b  outputs a control signal corresponding to transmission power (dB). A power supply control unit  15  converts the control signal into a signal corresponding to the control characteristic of the power supply. A D/A converter  16  converts the control signal from a digital signal to an analog signal. The voltage of the variable voltage power supply  3   a  is controlled by using the control signal obtained in this manner. The original purpose of this control is to apply, to the power amplifier  2 , a minimum necessary power supply voltage which can generate transmission power without any distortion so as to reduce the power consumption of the power amplifier  2 . Using this technique makes it possible to greatly reduce current at the time of low-power output. 
       FIG. 7  shows the arrangement of a transmitter having a distortion control function which is based on the transmitter in  FIG. 6 . The same reference numerals as in  FIGS. 1 and 6  denote the same parts in  FIG. 7 . In this exemplary embodiment, the waveform analyzing unit  13 , an adder  17 , and the power supply control unit  15  constitute a distortion control device  61 . The adder  17  adds a control signal corresponding to the transmission power (dB) output from the digital baseband unit  12  to the back-off value or MPR value output from the waveform analyzing unit  13 . The power supply control unit  15  then controls the variable voltage power supply  3   a  by using the obtained addition signal. 
       FIG. 8  is a graph showing the effect of this exemplary embodiment. Referring to  FIG. 8 , the abscissa represents the transmission power (dB) output from the digital baseband unit  12   b ; and the ordinate, the controlled power supply voltage. The solid line indicates a state in normal times. When the back-off value or MPR value output from the waveform analyzing unit  13  is added, control is performed in the manner indicated by the dotted line. In this manner, the power supply voltage of the power amplifier  2  can be increased by the necessary back-off indicated by the arrow. This can increase the current flowing to the power amplifier  2  and reduce the distortion of transmission power, thereby decreasing the adjacent channel leakage power. 
     A problem of this exemplary embodiment is that current increases. Unlike in the two exemplary embodiments described above, since no power reduction occurs, this exemplary embodiment is free from the demerit that the area of the cell decreases. In addition, since a DC/DC converter capable of raising/lowering a voltage is commercially available as the variable voltage power supply  3   a , it is possible to use this device. 
     Although the three exemplary embodiments described above can be executed singly, they can be executed in combination. That is, the exemplary embodiment shown in  FIG. 1  can be combined with the exemplary embodiment shown in  FIG. 7 , or the exemplary embodiment shown in  FIG. 4  can be combined with the exemplary embodiment shown in  FIG. 7 . If, for example, MPR values are obliged to be discretely and strictly set in increments of 1 dB or 0.5 dB, it is possible to decrease the transmission power by the method in  FIG. 1  or  4  using the discrete value and compensate for a deficient fractional portion by the method in  FIG. 7 . 
     The above exemplary embodiments can further comprise a function of calculating the estimated value of a back-off value from a combination of weighting relative values β of a plurality of code channels constituting a baseband signal.