Patent Publication Number: US-10330616-B2

Title: System and circuit for obtaining impedance or dielectric measurements of a material under test

Description:
TECHNICAL FIELD 
     The invention relates generally to systems, circuits and methods for determining characteristics of a material under test (MUT) using impedance or dielectric measurements of that MUT. 
     BACKGROUND 
     The use of impedance to measure the characteristics of construction, manufacturing, and biological materials by the application of impedance tomography and impedance spectroscopy is increasing. 
     The subject matter of U.S. Pat. No. 5,900,736, U.S. Pat. No. 6,414,497 and U.S. Pat. No. 7,219,024; US Patent Publication No. 2009/0270756; US Patent Publication No. 2012/0130212; US Patent No. 2013/0307564, Provisional U.S. Patent Application No. 61/703,488 (filed on Sep. 20, 2012); US Patent Publication No. 2014/0278300, US Patent Publication No. 2015/0137831; US Patent Publication No. 2015/0212026; Provisional U.S. Patent Application No. 62/039,204 (filed on Aug. 19, 2014); and Provisional U.S. Patent Application No. 62/103,835 (filed on Jan. 15, 2015) describe some impedance-related techniques for determining characteristics of materials, and are each incorporated by reference herein in its entirety. 
     SUMMARY 
     The system and circuit of the present subject matter relate to the measurement of the impedance of a MUT, as well as electronic devices and/or components for performing such measurements at a specific frequency or over a range of frequencies, with provisions for the self-adjustment of the transmit and reference signals to produce a measured signal within a desired range of the electronic measuring components over the frequency range, based upon the strength of the measured signal. The present subject matter provides an electronic circuit, a system, and a method to apply an electronic circuit which: 1) generates a transmit signal and a reference signal at a specific frequency or over a range of frequencies; 2) transmits a signal to a material under test (MUT) (which may include one or more sub-components); 3) compares the strength (and/or magnitude) of the transmitted signal passing through the MUT to the reference signal; 4) determines the phase relationship between the signal transmitted through the MUT relative to the reference signal; 5) computes the impedance or dielectric of the MUT (and in some cases, sub-components); and 6) applies the measured impedance or dielectric to characterize a physical property of the MUT. The approaches described herein can include characterization methods for the measuring circuit board and sensor system, as well as a method to gather data with the circuit board and sensor system. 
     Various embodiments of the disclosure relate generally to a system and circuit for the measurement of the impedance or dielectric of a material under test (MUT). In some cases, the system includes a circuit having level detectors to measure the change in strength between a reference signal and a transmit/receive (return) signal having passed through the MUT. The system can include at least one computing device configured to evaluate the measured signal levels and adjust those signals within range of the level detectors and other circuit components. Circuits according to various embodiments can include a phase determiner for determining the phase shift between the reference signal and the receive (return) signal that passes through the MUT. According to various embodiments, the measured difference in signal strength and phase are used to compute the complex impedance (or dielectric properties) of the MUT. This impedance or dielectric property can be correlated with a physical property of the MUT. The system may be operated at a single frequency, or over a range of frequencies. 
     In some particular embodiments, a system can include: a signal generator; a transmitting electrode connected with the signal generator and in electromagnetic communication with a material under test (MUT); a receiving electrode connected with the signal generator and in electromagnetic communication with the material under test (MUT); a reference level detector connected with the signal generator in parallel with the transmitting electrode; an absolute level detector and a phase determiner connected with the receiving electrode; a phase determiner connected with the receiving electrode and the signal generator; and at least one computing device connected with the signal generator, the phase determiner, the reference level detector, and the absolute level detector, the at least one computing device configured to: send a control signal to the signal generator to initiate: an transmit excitation signal to the MUT via the transmitting electrode at a selected frequency, and a reference signal to the reference level detector; receive a reference level signal from the reference level detector; receive a return signal from the MUT via the receiving electrode and the absolute level detector; receive a phase signal from the phase determiner; and record the reference level signal, the phase signal and the return signal. 
     In some particular embodiments, the phase determiner includes a means to generate a comparison signal at the same frequency as the reference and transmitting frequency but with varying phase shifts relative to reference and transmit signals. This comparison signal is superimposed on or summed with the receive signal. The phase of the comparison signal which, when superimposed on the receive signal, produces the peak voltage defines the value phase shift. 
     In one embodiment, the phase determiner uses a time-of-flight method with a Time-to-Digital Conversion (TDC) chip to directly measure the time between edges on the comparison and receive (return) signals. This measured time may be used to compute the phase shift between the two signals. 
     In some cases, a system can include: an electromagnetic signal generator operating at a fixed frequency or over a range of frequencies generating two parallel signals, a Reference Signal and a Transmit Signal; a transmitting electrode connected with the Transmit Signal from the signal generator and in electromagnetic communication with a material under test (MUT); a receiving electrode connected in electromagnetic communication with the material under test (MUT) generates a Receive Signal; a reference absolute level detector connected with the Reference Signal from the signal generator; a receive absolute level detector connected with the Receive Signal from the receiving electrode; a time-of-flight phase determiner means connected with the Receive Signal from the receiving electrode and the Reference Signal from the signal generator; and at least one computing device connected with the signal generator, the phase determiner, the reference absolute level detector, and the receive absolute level detector, the at least one computing device configured to: send a control signal to the signal generator to initiate: an excitation signal to the MUT via the transmitting electrode at a selected frequency or over a range of frequencies, and a reference signal to the reference absolute level detector; receive a reference level signal from the reference absolute level detector; receive a return Receive Signal from the MUT via the receiving electrode and the absolute level detector; receive a phase signal from the phase determiner; and record the Reference Signal level, the phase signal, and the Receive Signal level. 
     In certain cases, the at least one computing device is further configured to use the signal levels and phase angle to compute either the impedance or dielectric of the MUT to determine a physical characteristic of the MUT. 
     In particular embodiments, the physical characteristic includes at least one of density, water content, or physical composition. 
     In some embodiments, the system further includes a set of circuit board terminals connecting the circuit board with the transmitting electrode and the receiving electrode; and a calibration circuit conductively coupled with the circuit board terminals, the calibration circuit configured to short circuit across the circuit board terminals, wherein the at least one computing device is further configured to: generate a calibration signal across the calibration circuit; receive and compare the calibration signal absolute level reading to the Reference Signal absolute level reading; and modify the control signal to the signal generator in response to the calibration signal deviating from the reference level signal by greater than a threshold. 
     In certain embodiments, the threshold is equal to approximately a one percent deviation. 
     In particular embodiments modifying the control signal to the transmit amplifier includes providing instructions to adjust the excitation signal to the transmitting electrode by an amount corresponding with the deviation between the calibration signal and the Reference Signal. 
     In some cases, the system further includes a transmit amplifier connected with and located between the signal generator and the transmitting electrode, the transmit amplifier for amplifying the excitation signal prior to transmission into the MUT. 
     In certain instances, the system further includes a fixed level attenuator connected with and located between the signal generator and the reference absolute level detector, wherein the fixed level attenuator is configured to decrease in amplitude of the Reference Signal to approach an amplitude of the Receive Signal. 
     In particular embodiments, the system further includes: a set of circuit board terminals connecting the circuit board with the transmitting electrode and the receiving electrode; and a calibration circuit conductively coupled with the circuit board terminals, the calibration circuit configured to apply at least one of a resistive or a capacitive load across the set of circuit board terminals, wherein the at least one computing device is further configured to: generate a calibration signal across the calibration circuit; receive and compare the calibration signal absolute level reading to the Reference Signal absolute level reading; and modify the control signal to the signal generator in response to the calibration signal deviating from the reference level signal by greater than a threshold. 
     In certain cases, applying the at least one of the resistive or the capacitive load includes applying both the resistive load and the capacitive load across the set of circuit board terminals, wherein the resistive load and the capacitive load are selected based upon an emulated impedance of the MUT. 
     In some embodiments, the at least one computing device is further configured to: iteratively send a control signal to the transmit amplifier to adjust a gain on the transmit amplifier until the calibration signal deviates from the reference absolute level signal by less than or equal to the threshold. 
     In certain implementations, the at least one computing device is further configured to: store an amplification value corresponding with the calibration signal deviating from the reference signal by less than or equal to the threshold. 
     In particular cases, the at least one computing device is further configured to: store a corresponding phase angle for the reference signal and the receive signal at the amplification value, received from the phase determiner. 
     In some embodiments, the system further includes an MUT calibration circuit electromagnetically coupled with the transmit electrode and the receive electrode, the MUT calibration circuit configured to emulate an impedance response of the MUT at the receive electrode, wherein the at least one computing device is further configured to: generate a calibration signal across the calibration circuit; receive and compare the calibration signal absolute level reading to the Reference Signal absolute level reading; and modify the control signal to the signal generator in response to the calibration signal deviating from the reference level signal by greater than a threshold. 
     In certain instances, the calibration circuit includes an equivalent electrode array configured to emulate the impedance response of the MUT that is electromagnetically coupled to the transmit electrode and the receive electrode. 
     In particular embodiments, the at least one computing device is further configured to: iteratively send a control signal to the transmit amplifier to adjust a gain on the transmit amplifier until the calibration signal deviates from the reference absolute level signal by less than or equal to the threshold. 
     In some cases, the at least one computing device is further configured to: store a transmit amplification value corresponding with the calibration signal deviating from the Reference Signal by less than or equal to the threshold. 
     In certain implementations, the at least one computing device is further configured to: store a corresponding phase angle between the Reference Signal and the Receive Signal at the amplification value, received from the phase determiner. 
     In particular cases, the at least one computing device is further configured to: iteratively send a control signal to the phase determiner amplifier to adjust a gain on the amplifier until the receive signal and the reference signal levels are within an operating range of the edge detector of the phase determiner. 
     In some embodiments, the at least one computing device is further configured to: store a phase angle of the control signal corresponding with the return signal and the reference level signal being within the operating range of the phase determiner. 
     In certain implementations, the signal generator includes a direct digital synthesizer and a signal conditioner. 
     In particular instances, the absolute level detector provides an analog value of an absolute magnitude of a voltage of the reference signal to the at least one computing device. 
     In some cases, the system further includes a time-of-flight phase determiner means connecting the Reference Signal and Receive Signal to an edge detector which converts the sinusoidal signals to a square waves which are then processed by a Time-to-Digital Conversion chip which provides a precision measurement of the time between the edges of the two waves. 
     In particular cases, the at least one computing device is further configured to compute the phase angle between the Reference Signal and the Receive Signal by the following equation:
 
Phase Angle=360×(Phase Time/Cycle Time).
 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         FIG. 1  shows a general configuration of a sensor system including an impedance measurement circuit according to various embodiments of the disclosure. 
         FIG. 2  shows an example graphical depiction of the signal differences of reference signals and return signals from the sensor system of  FIG. 1  according to various embodiments of the disclosure. 
         FIG. 3  shows a particular configuration of the sensor system of  FIG. 1  according to various embodiments of the disclosure. 
         FIG. 4  shows a calibration system according to various embodiments of the disclosure. 
         FIG. 5  illustrates several equivalent circuit models compatible with the calibration system of  FIG. 4 . 
         FIG. 6  shows a flow diagram illustrating processes in a method according to various embodiments of the disclosure. 
         FIG. 7  shows an equivalent circuit model for the calibration system  220  of  FIG. 4  according to various embodiments of the disclosure. 
         FIG. 8  shows an equivalent circuit model for the calibration system  220  of  FIG. 4  according to various embodiments of the disclosure. 
         FIG. 9  shows a sensor system including a calibration fixture at the sensor system level according to various embodiments of the disclosure. 
         FIG. 10  is a schematic depiction of sensor calibration fixtures and equivalent circuit models for the system of  FIG. 9 , according to various embodiments of the disclosure. 
         FIG. 11  shows a flow diagram illustrating a method according to various embodiments of the disclosure. 
         FIG. 12  shows a system according to various embodiments of the disclosure. 
         FIG. 13  shows a flow diagram illustrating a method according to various embodiments of the disclosure. 
         FIG. 14  illustrates alternate circuit embodiments for terminations in the system of  FIG. 12 , according to various embodiments of the disclosure. 
         FIG. 15  illustrates a termination circuit configuration according to various embodiments of the disclosure. 
         FIG. 16  illustrates a phase sweep determiner system according to various embodiments. 
         FIG. 17  is a graphical depiction of superimposed comparison signals and receive (return) signals versus phase angle shown with reference to the system of  FIG. 16 . 
         FIG. 18  is a graphical depiction of percent error versus phase angle shown with reference to the system of  FIG. 16 . 
         FIG. 19  shows a flow diagram illustrating a method of performing phase determination according to embodiments of the disclosure. 
         FIG. 20  illustrates a time-of-flight phase determination system according to various embodiments of the disclosure. 
         FIG. 21  illustrates an embodiment of methodology phase determination system according to various embodiments of the disclosure. 
         FIG. 22  shows a flow diagram illustrating a method for determining a phase angle according to various embodiments of the disclosure. 
         FIG. 23  shows a sensor system according to various embodiments of the disclosure. 
     
    
    
     DETAILED DESCRIPTION 
     The various methods and procedures described here are related to the determination of the impedance characteristics of a material under test (MUT) at a single selected frequency. A single frequency is adequate for tomographic analysis as described in US Patent Publication Nos. 2010/037361, 20130307564, and 2015/0137831, and U.S. Patent Application No. 61/703,488; and or for the determination of physical properties of selected materials that act as pure capacitors, such as hot mix asphalt, as described in U.S. Pat. Nos. 5,900,736 and 6,414,497. Each of these applications, publications and issued patents are hereby incorporated by reference in its entirety. 
     An illustrative schematic view of a system  100  according to various embodiments is illustrated in  FIG. 1 . The MUT  10  is in electromagnetic communication with an array of electrodes,  109  and  110 . The electromagnetic communication may be either electrically conductive or electrically non-conductive, and in some cases electrodes  109 ,  110  are separated from a surface of MUT  10  by a distance (or, off-set). In the following discussions the non-conductive application will be discussed in relatively greater detail. According to various embodiments, an electromagnetic signal is generated by a signal generator  101 , triggered by a control signal  113  from a microprocessor  106 . In many cases, two signals are transmitted from signal generator  101 . One signal is a reference signal  103 , which is transmitted to a level detector  104 , the output of which is transmitted as an analog voltage level signal to microprocessor  106 . The microprocessor  106  selected can have a 12-bit A-to-D input port. The other signal, excitation signal (TX 102 ) or “transmit” signal, is transmitted to a circuit board terminal  107 , on a circuit board that is electrically coupled with the transmitting (sensor) electrode  109 . After passing through MUT  10 , the transmitted signal is received by the receiving (sensor) electrode  110 , which is electrically coupled with the circuit board through terminal(s)  107 . From the circuit board terminals  107 , a receive signal  112  is transmitted to a level detector  104 , the output of which is transmitted as an analog voltage level signal to microprocessor  106 . In various embodiments, the levels of the reference signals  103  and receive signals  112  can be recorded by microprocessor  106  and stored for subsequent use/combination with the phase angle between the reference signal  103  and receive signal  112 , in order to compute the impedance and/or dielectric value of the MUT  10 . This then can be correlated with a desired physical property of the MUT  10  (such as density for asphalt). A circuit board level calibration component  120  is also shown, and is discussed further herein. 
     Basic quantities measured at a given frequency can include the change in the magnitude between the reference signal (level)  103  and the received signal (level)  112 , due, for example, to the resistive dissipation of the signal as it passes through MUT  10 ; and the phase shift of the received signal  112  relative to the reference signal  103 , due to, for example, capacitive effects of the MUT  10 . In general, physical materials may not produce detectable inductive losses without specifically setting tests to induce such losses (which may be feasible with only certain classes of materials).  FIG. 2  illustrates the effects passing reference signal  103  through MUT  10  has on received signal  112 . The differences between reference signal  103  and receive signal  112  are discussed further herein. 
     While the system(s) disclosed herein may include various conventional components, as well as configurations specific to the aspects of the disclosure, it is understood that components may be included in the design of the circuit to achieve different operational characteristics without affecting the scope of the operation of the system(s). 
     Turning to  FIG. 3 , an additional system  200  according to various embodiments is shown schematically. Common elements between the figures can represent substantially common components. As shown in  FIG. 3 , system  200  can include a signal generator  101  which can include a single channel Direct Digital Synthesizer (DDS) (such as the Analog Devices AD 9911)  134 , in some embodiments. In various embodiments, coupled with the DDS  134  is a signal conditioner  135  configured to condition the control signal  113  prior to splitting that signal between a reference signal  103 , and an excitation signal  102 . In some embodiment, reference signal  103  is passed through a specified fixed level attenuator  108 , and then split again with one leg going to a phase determiner  140 , and another leg going to an absolute level detector  104  (e.g., an AD8310) with a reference termination  130 , known in the art. Absolute level detector  104  can produce an analog value of the absolute magnitude of the voltage of reference signal  103 . This analog value may act as an input to microprocessor  106 , e.g., at a 12-bit A-to-D port (e.g., a Microchip dsPIC33EP512GP). 
     The second leg of the signal from signal conditioner  135  passes through an amplifier,  137  as excitation signal  102 , to a terminal  107  on the circuit board, connected to a transmit (sensor) electrode,  109 . The transmit electrode  109  and the receive electrode,  110 , are configured to electromagnetically communicate with MUT  10 . Electromagnetic communication between the electrodes  109 ,  110  and MUT  10  may be either electrically conducting or electrically non-conducting, as shown in  FIG. 3 . Receive electrode  110  is connected to circuit board terminal  107 , and receives a response signal (receive signal  112 ) from MUT  10 . From the terminal  107 , the receive signal  112  is split between a leg going to the phase determiner  140 , and an absolute level detector  104 , with a Reference Termination  111 , known in the art. Absolute level detector produces an analog value of the absolute magnitude of the voltage of the reference signal  112 . This analog value may act as another input to microprocessor  106 , e.g., at a second 12-bit A-to-D port. At this point, reference signal  103  and the receive signal  112  received at phase determiner  140  is passed as a digital value of the time between trigger points of the reference signal  103  and receive signal  112 , shown as phase  117 . This digital time value of phase  117  can be inputted to microprocessor  106 , and stored in some cases. As shown DDS  134  can be controlled by microprocessor  106 , e.g., through control signal  113 , by specifying the frequency and amplitude of the signal generated by DDS  134 . 
     In various embodiments the fixed level attenuator  108  can minimize the amount that the TX Amplifier  137  needs to amplify the excitation signal  102  as it passes through MUT  10 , so that both receive signal  112  and attenuated reference signal  103  are within the dynamic range of the absolute level detectors  104 . 
     In various embodiments, in order for the system to calibrate the circuit board, the voltage difference between the reference signal  103  and receive signal  112  are eliminated, without MUT  10  at the board level and the sensor system level. This is accomplished by using a “short” for the board calibration  120  (as illustrated in  FIG. 4 ), and adjusting the TX amplifier  137 , so that the RX Level  116 , equals the REF Level  115 . Also, in some cases, the phase difference between reference signal  103  and receive signal  112  are known at the board level and the sensor system level. Approaches for standardizing or calibrating system  200  are described with respect to calibration system  220 , illustrated schematically in  FIG. 4 . As compared with system  200 , calibration system  220  further includes a board calibrator  120  located between the terminals  107  of the circuit board.  FIG. 5  illustrates further schematic details of board calibrator  120 , in several calibration scenarios. A first configuration, as shown, is to “short” the two electrodes. By “shorting,” a conductor is placed between the electrodes  107 . However, the use of a “short” with this configuration of the board circuit will not be effective due to the presence of attenuator  108  ( FIG. 4 ). In order to characterize the circuit board, a resistive or capacitive load is applied as shown in  FIG. 5 . Another approach is to use a combination of resistive and capacitive loads (as shown in  FIG. 5 ) that emulate the approximate impedance characteristic of the MUT. In general, the range of the expected impedance characteristics of the MUT  10  at a given frequency is known, as this would be a requirement for the effective design of the sensor circuit. For examples, if the MUT  10  is hot mix asphalt, it is known in the art that this material acts as a pure capacitor with a dielectric value in the range of 4 to 7, depending on the degree of compaction. Soils used in construction also have a defined range of dielectric depending on soil type, moisture level, and degree of compaction. Therefore, using a Resistor-Capacitor (RC) circuit within the range of the impedance characteristics of the MUT (e.g. asphalt), or a series of RC circuits that cover the range of the impedance characteristics of the MUT (e.g. soil), can provide a characterization of the variations of the individual components of the system  200  ( FIG. 3 ) and their distortions of the signals in the system  200  as a whole.  FIG. 6  presents a characterization or calibration procedure for system  200  ( FIG. 3 ). Referring to calibration system  220  in  FIG. 4  and the process in  FIG. 6  collectively, processes can include: 
     P 100 : Select a transmission frequency at electrodes (transmit electrode(s)  109 ) specific to the type of MUT  10  being tested with the configuration of system  200  (for example, in general, a frequency in the range of 10 MHz to 20 MHz is acceptable with materials encountered in civil construction projects); 
     P 101 : Place the selected board calibrator  120  (see  FIG. 5 ), appropriate simulated impedance(s) for the specific MUT  10  across terminals  107 , and initiate the control signal  113  to DDS  134 , to generate the signal at the selected frequency and voltage level; 
     P 102 : Adjust the gain on amplifier  137  (via control signal  113  from microprocessor  106 ); 
     P 103 : If the RX Level  116  does not equal the REF level  115 , adjust the amplifier  137  accordingly until the RX level  116  and REF level  115  are identical; 
     P 104 : Store the value of the amplification from amplifier  137  at microprocessor  106 , UCI  200  or other data storage device; 
     P 105 : Determine the phase angle between the REF level  115  and RX signal  116  according to the selected phase determiner  140 ; and 
     P 106 : Record and store the measured phase angle from phase determiner  140 , e.g., at microprocessor  106 , UCI  200  or other data storage device. Various aspects of the phase determiner  140  are discussed in detail further herein. 
     Having characterized/calibrated system  200 , sensors (electrodes  109 ,  110 ) and their associated connectors, as well as an air gap or an insulator layer (optional) between electrodes  109 ,  110  and MUT  10 , e.g., for a non-conducting electromagnetic communication with MUT  10 , may be added. Additionally, after characterizing/calibrating system  200 , impedance characteristics of the MUT  10  can be calculated, e.g., using an equivalent circuit model with a combination of resistors and capacitors in series and/or parallel arrangements. That is, electrodes  109 ,  110  can be used to obtain data used to calculate a complex impedance, Z, of each component in the measured structure. This is illustrated schematically in  FIG. 7 . The circuit board is configured to measure ZM, which is a function of the impedances of all of the individual components of the measured structure (e.g., terminals  107 , electrodes  109 ,  110 , air gap and MUT  10 ):
 
 ZM=f ( Z 1, Z 2, Z 3, Z 4, Z 5, Z 6, Z 7)
 
A target measurement for MUT  10  is Z4. That is, an objective of the characterization procedure outlined in  FIG. 6  is to assure that the effects of the impedances Z1 and Z7 at the terminals  107 , which include the effects of all the measured structure components and the circuit design of the board, can be ignored in determining the value of Z4, attributable to MUT  10 . However, it still may be difficult to transition from the measured impedance at terminals  107  to the impedance of MUT  10 . If the characterization or calibration is performed at the level where the electrodes are placed into electromagnetic communication with the MUT  10 , the point of the measured impedance is now moved to the electrode/air-insulator (or electrode  109 , 110 /MUT  10 ) interface. This simplified scenario is shown schematically in  FIG. 8  where the sensor characterization now occurs at the sensor electrodes  109  and  110 . In this case, the measured impedance, ZMS, is just the sum of the impedance of the two air gaps between electrodes  109 ,  110  and MUT  10  (or another insulator), and the MUT  10 :
 
 ZMS=Z 3+ Z 4+ Z 5
 
Solving for Z4 becomes much simpler in this approach when compared with the configuration in  FIG. 7 .
 
     The characterization or calibration of the system  200  may be performed in a similar manner as described with respect to the circuit board herein.  FIGS. 8 and 9  illustrate components in an additional calibration system  400 , which utilizes a sensor calibrator  121  being placed across the sensor electrodes  109  and  110 . The equivalent RC circuit of the sensor calibrator  121  is similar to that used to emulate the impedance characteristics of MUT  10  with board calibrator  120  shown in  FIG. 5 . However, as illustrated in  FIG. 10 , rather than having a wire contact to the electrodes  109 ,  110 , an equivalent electrode array  500  is used to provide electromagnetic wave-based communication that better characterizes the electromagnetic communication between electrodes  109 ,  110  and MUT  10 . In  FIG. 10 , circular (e.g., copper) electrodes are shown as examples of electrodes  109 ,  110 , and may be mounted on a support structure, typically a circuit board material such as FR4 or G9 (glass reinforced epoxy laminate sheets). Other electrode shapes and materials are also possible, and other support structure materials are also possible. According to various embodiments, sensor calibrator  121  includes the sensor characterization electrodes  509 ,  510 , which oppose electrodes  109 ,  110 . As described herein, electrodes  109 ,  110  are connected to the board terminals  107 . The sensor characterization electrodes  509 ,  510  are connected to an MUT impedance emulation circuit  520 , which is configured to emulate the impedance of one or more particular types of MUT  10 . 
     An active version of an impedance emulator is presented in US Patent Publication No. 2014/0278300. This device varies the emulated impedance over a range of frequencies. 
       FIG. 11  presents a flow diagram illustrating a standardization/calibration process which may be used in conjunction with system  400  ( FIG. 9 ), including equivalent electrode array  500  ( FIG. 10 ). Referring to  FIGS. 9-11 , the process may include the following sub-processes: 
     P 200 : Select a frequency specific for the MUT  10  being tested with the selected circuit board configuration (e.g., for tomographic measurements, frequencies in the range of 10 MHz to 20 MHz are generally acceptable; e.g., for spectrographic measurements, such as for soils, frequencies in the range of 10 MHz to 50 MHz are used as described in U.S. Pat. No. 7,219,024); 
     P 201 : Place the selected sensor calibration fixture  121  ( FIG. 9 ), appropriate for the simulated impedance(s) of the specific MUT  10 , in electromagnetic communication with electrodes  109 ,  110  and initiate the control signal  113  to DDS  134 , to generate excitation signal  102  at the selected frequency and voltage level determined in process P 200 ; 
     P 202 : Adjust the gain on amplifier  137  ( FIG. 9 ) (via control signal  134  from microprocessor  106 ); 
     P 203 : If the RX level  116 , does not equal REF Level  115 , adjust amplifier  137  accordingly until the RX level  116  and REF level  115  are approximately (+/−one percent) identical; 
     P 204 : Store the value of the amplification from amplifier  137  at microprocessor  106 , UCI  200  or other data storage device; 
     P 205 : Determine the phase angle between the REF level signal  115  and RX level signal  116  according to the selected phase determiner  140 ; and 
     P 206 : Record and store the measured phase angle  117  at microprocessor  106 , UCI  200  or other data storage device. 
     At this point, according to various embodiments, the sensor system  600  shown in  FIG. 12  may be configured to measure the impedance of MUT  10 . System  600  can include various components shown and described with respect to systems  100 ,  200 ,  300  and  400 , discussed herein, and can include: (i) Microprocessor  106 ; (ii) Signal Generator/Analyzer  302  connected with microprocessor  106 ; (iii) electrodes  109 ,  110  and their related array/assembly  301 , connected with signal generator/analyzer  302 ; and (iv) MUT  10 . As shown in  FIG. 13 , one process for measuring the impedance of MUT  10  using system  600  can include the following sub-processes: 
     P 300 : Select the frequency specific to the MUT  10  being tested with the configuration of sensors  109 ,  110 , and set the amplifier  137  and phase angle as determined by either process  100  or process  200 ; 
     P 301 : Place MUT  10 , in electromagnetic communication with sensor electrodes  109 ,  110 ; 
     P 302 : Measure and record the REF level  115  and RX level  116  at microprocessor  106 ; 
     P 303 : Determine whether RX level  116  and REF level  115  are in the operating frequency range of the electronic components used for the phase determination (see  FIG. 16 ). 
     P 304 : Adjust the gain on amplifier  137  until the RX level  116  and REF level  115  are within the operating frequency range of the electronic components used for phase determination  140  ( FIG. 16 ); 
     P 305 : Record the phase angle  117 , at microprocessor  106 , UCI  200  or other data storage device; 
     P 306 : Determine the impedance/dielectric of the MUT as described US Patent Publication Nos. 2013/0307564 and 2016/0161624, each of which is incorporated by reference herein; and 
     P 307 : Apply the conversion algorithm to determine the desired physical characteristic of the MUT as described US Publication Nos. 2013/0307564 and 2016/0161624. 
     Referring back to  FIG. 3 , and with reference to  FIG. 14 , the reference termination  130  and the receive termination  111 , can be defined using termination functions shown in  FIG. 14 . Alternate A is a simple fixed resistor. Alternate B is a fixed resistor-capacitor circuit. However, according to various embodiments, the circuit  700  shown in  FIG. 15  may be used as a termination circuit, incorporating both reference termination  130  and receive termination  111  functions. This termination circuit  700  can be particularly beneficial for the AD8310 version of absolute level detector  104 , however, if another model of absolute level detector  104  is used, this circuit  700  or a different termination circuit may be used. 
     Referring again back to  FIG. 3 , the phase determination may be accomplished by the phase sweep method discussed herein, or the time-of-flight measurement method, also discussed herein. The shortcomings of the phase sweep method will be discussed to illustrate why the time-of-flight method using a Time-to-Digital Conversion (TDC) chip is beneficial in the systems and approaches of the present disclosure. 
     Referring to  FIG. 16 , a system  800  is shown, where system  800  is configured to perform a phase sweep method for determining the phase angle shift between two signals according to various embodiments. This embodiment uses a dual channel sine wave generator  801  to generate a set of signals, as described herein. An alternate embodiment could use two single channel sine wave generators in place of the dual channel sine wave generator  801 . Dual channel sine wave generator  801  generates a signal that is divided, by a current-to-voltage converter  803 , into a reference signal  804  and a transmit signal  107 . Reference signal  804  is passed through a fixed attenuator  805 , and then to an absolute level detector  812 , as discussed with respect to various other systems herein. The output of absolute level detector  812  communicated to the microprocessor  814 . Transmit signal  807  can be passed through an amplifier  806  before being transmitted through the MUT  10 . The receive (return) signal  840  from MUT  10  is split, with a portion sent through a controllable amplifier (not shown, and under control of the microprocessor  814 ) and then to an absolute level detector  813 , the output of which is communicated to a microprocessor  814 . Meanwhile, reference signal  804  is transmitted through a distinct absolute level detector  812 , and then to microprocessor  814 . The magnitude variation between the two signals, reference  804  and receive (return), is determined and stored for use in the computation of the measure impedance of MUT  10 , as described herein. The receive (return) signal  840  from MUT  10  is also transmitted to a summing amplifier  815 , which is part of a phase determiner circuit. Turning back to the sine wave signal generator  801 , a comparison signal  817  is generated by dual channel sine wave signal generator  801  and converted using current-to-voltage converter  802 . The frequency of comparison signal  817  and (reference) transmit signal  807  are set to be identical. The relative phase angle between the two signals is varied. The size of the phase increment, e.g., one degree, can be selected by the user or designer of the system  800 . In general, the range over which the phase angle between the transmit signal  807  and comparison signal  817  is shifted is ±180 degrees. In practice, however, either prior data with the system  800  and MUT  10  or an a priori analysis may provide an estimate of the expected phase angle shift between transmit signal  807  and comparison signal  817 . Using this base value of the expected phase angle shift as the center of the range to be swept, a swept range of, e.g., ±20 degrees would be adequate to determine the actual phase angle of a particular measurement. 
     Returning to the path of comparison signal  817 , after current-to-voltage conversion  802 , it is passed through a variable attenuator  818 , which is controlled by microprocessor  814 . From the variable attenuator  818 , it is transmitted to an absolute level detector  820 , the output of which is communicated to microprocessor  814 . After leaving absolute level detector  820 , the signal is joined with the receive (return) signal  840  from MUT  10  in summing amplifier  815 . The sum of the two signals is passed to a peak detector  816 , the output of which is communicated to microprocessor  814 . 
     The signal from absolute level detector  820  is compared to that from the absolute level detector  813  at microprocessor  814 , and used, via control feedback signal  830  to control variable attenuator  818  to modify the output signal from variable attenuator  818  to match the output signal from absolute level detector  813 . When the phases of the comparison signal  817  and receive (return) signal  840  from MUT  10  are in phase, the amplification value of summing amplifier  815  will be equal to twice that of the receive (return) signal  840 . When the phases of comparison signal  817  and receive (return) signal  840  from MUT  10  are ±90 out of phase, the amplification value of summing amplifier  815  is one; when ±180 out of phase, the value is zero. This is illustrated in the graphical depiction of the superimposed signals (comparison  817  and receive  840 ) v. phase angle of  FIG. 17 , where the superposition of comparison signal  817  with varying phase angles between receive signal  840  and the comparison signal  817  is made on receive signal  840 . 
     One problem with the phase sweep approach is the amount of time required to sweep through the range of phase angles. In the most general (worse) case, there are 360 phase change readings. The time required to sweep through all 360 angles and record the data from summing amplifier  815  and peak detector  816  is about 400 milliseconds. This would not be as problematic if the measurement device is stationery, and measures the same volume of MUT  10 . However, systems according to various embodiments disclosed herein are designed to allow operation on a mobile platform, e.g., as described in WO Publication No. 2016/115318 (incorporated by reference in its entirety). When a sensor system is mounted on a non-stationary component, e.g., on a pavement roller traveling several (e.g., 3) miles per hour, the sensor will move a certain distance, over time (e.g., about 1.8 feet in the 0.4 seconds), while the phase angle sweep is proceeding. In some cases, the sensor moves a plurality of feet, over fractions of a second, before the entirety of the phase angles can be swept. This does not include additional time to record and process the phase sweep data. In these scenarios, there is no guaranty that the volume of the MUT  10  being measured throughout this time (and across this distance) remains constant. 
     Another problem with the phase sweep approach relates to the precision with which the peak of the phase sweep can be identified. The characteristic of summing two sine waves is that at the peak, the signal becomes flat. This is illustrated in the graphical depiction of the phase sweep in  FIG. 17  based upon system  800 . This graphical characteristic illustrates the principle that as the peak is approached, the amount of signal change with the phase angle becomes smaller. This is illustrated in  FIG. 18 , which shows a graphical depiction of percent error for comparison signal  817  versus phase angle error. As shown, an error of less ±0.8% in the reading of the summed signal can result in an error in the determination of the correct phase angle between the two signals of up to ±10 degrees. 
     These and other problems with the phase sweep phase approach make it embodiment particularly challenging approach to implement in a moving system. 
       FIG. 19  shows a flow diagram for system  800  ( FIG. 16 ) illustrating a method for determining a phase angle difference between receive signal  840  and comparison signal  817  according to various embodiments of the disclosure. This flow diagram is discussed with continuing reference to  FIGS. 1, 3, 4, 9, 12, 16, 20, and 21 , but may be particularly applicable to system  800  in  FIG. 16 . As shown, the process can include: 
     Process P 400 : Generate a comparison signal  817  in signal generator  801  with the same frequency as reference signal  804  and transmit signal  807 , and at the same phase angle. 
     Process P 401 : Obtain the readings for the receive Signal  840  from absolute level detector  813  and for the comparison signal from absolute level detector  820  from microprocessor  814 . 
     Process P 402 : Compare the values of readings from absolute level detector  813  and absolute level detector  820  to determine if they are equal. If yes, proceed to P 404 ; if not, proceed to P 403 . 
     Process P 403 : Send a control feedback signal  830  to variable attenuator  818  to adjust the level of comparison signal  817  so that the readings of absolute level detector  813  and absolute level detector  820  are equal. 
     Process P 404 : Generate a series of comparison signals  817  in signal generator  801  with the same frequency as reference signal  804  and transmit signal  807 , but varying the phase angle ±180 degrees. The resulting comparison signal  817  from the phase sweep process are directed along with receive signal  840  to the summing amplifier  815 . 
     Process P 405 : Searching for the peak in the phase sweep, which may be accomplished in a number of different ways. As shown in  FIG. 16 , a peak detection circuit  816  may be used to detect the peak in the phase sweep. Alternately, a computational method may be used in which a series of moving averages may be computed and compared. However, as shown in  FIG. 18  and discussed herein, the variation in signal strength as the phase angle is matched is very small. As such, small errors in the determination of the superimposed signal strength can result in large errors of the selected phase angle. 
     Process P 406 : Conduct a test to determine if the peak value of summing amplifier  815  has been determined. If yes, proceed to P 408 ; if no, proceed to P 407 . 
     Process P 407 : Continue to search for the peak across the phase sweep if the result of P 406  is negative. 
     Process P 408 : Record the identified value of the phase angle difference between comparison signal  817  and receive signal  840  in the microprocessor  814  for the computation of the impedance or dielectric of MUT  10 . 
     Turning now to  FIG. 20 , a system  900  for computing phase using a time-of-flight approach according to various embodiments is shown. System  900  may have some similar components as system  200  shown in  FIG. 3 . System  900  can additionally include a phase determiner  140 , including time-of-flight circuit components such as a variable gain amplifier  141 , an edge detector  142 , and a time-of-flight precision timer  143 . According to various embodiments of the disclosure phase determiner  140  enhances the phase shift calculation compared with conventional systems, and allows system  900  to precisely calculate phase shift between signals, as described herein. 
     An example of phase determiner  140  using the time-of-flight approach is shown according to various embodiments in  FIG. 21 . The time-of-flight circuit (system  900 ) includes phase determiner  140 , which may include a precision timer  143 . Precision timer  143  may provide a digital reading of the time between two triggers. In some cases, phase determiner  140  is designed to provide a response similar to a Time-to-Digital Conversion chip (e.g., Texas Instruments TDC7200), however, other conventional models could be used as well. In order to provide a proper trigger signal, the sinusoidal reference signal  103  and receive signal  112  are converted into a square wave. This can be accomplished, for example, by using an edge detector  142  (e.g., the Analog Devices AD8612 chip or another conventional model). However, in the example of AD8612 as edge detector  142 , this chip has a limited input dynamic range. In this example, in order to assure that the incoming signals are within the dynamic range of the AD8612 chip, the signals are passed through a dual channel variable gain amplifier  141 , which may include an ADRF6510 amplifier in some example embodiments. As shown in  FIG. 21 , the edge detector  142  chip can convert the sinusoidal ref signal  103  and receive signal  112  to square waves, which can be transmitted to the precision timer  143 , which provides a digital reading of the time between the zero crossings of the two signals. This digital time is then transmitted to the microprocessor  106 . Because the quality of the signal passing through the edge detector  142  is dependent on the incoming signals being within the dynamic voltage range of the chip, the timing detected by the precision timer  143  may be unstable. In these cases, microprocessor  106  can evaluate the phase time signal  117  from the precision timer  143  to determine whether that phase time signal  117  is stable, and if not, microprocessor  106  instructs variable gain amplifier  141  (via gain control signal  149 ) to modify one of its gain settings. Microprocessor  106  iterates this process until phase time signal  117  received from precision timer  143  is stable. The computation of the phase angle between reference signal  103  and receive signal  112  is based on the time measured by the phase determiner  140 , as described above, and the cycle time of the signals. The cycle time (or, period) of a signal is equal to the inverse of its frequency. For example, the frequency of the excitation signal  102  used to determine the density of hot mix asphalt is 13.6 MHz. In one example, the DDS  134  (e.g., AD9911 chip) has a frequency accuracy of at least 0.1 Hz. Thus, in the example of hot mix asphalt, the time for one cycle of the signal is 73.52×10 −9  seconds (or 73.52 nanoseconds). In another example, the precision timer  143  (e.g., TDC7200 chip) has a specification sheet standard deviation of 35×10 −12  seconds, and a resolution of 55×10 −12  seconds (or 55 picoseconds). As the resolution in this precision timer  143  is larger than the standard deviation, computation of the phase angle requires additional calculation. More specifically, because the chip measures two times (at the time from trigger for each wave), the potential resolution error of the phase time is 78×10 −12  seconds, which is equal to the square root of the sum of the squares of the resolution (standard deviation). In these cases, in order to compute the phase angle, the following equation may be used:
 
Phase Angle=360×(Phase Time/Cycle Time)
 
The potential error with this method (e.g., one standard deviation/resolution) at an example frequency of 13.6 MHz is 0.337 degrees.
 
     The above-noted approaches may have an inherently better precision than conventional approaches, e.g., the phase sweep approach described in US Patent Publication 2014/0266268 (also incorporated by reference herein in its entirety), and may also reduce cycle time for measuring phase angles relative to those conventional methods. In some cases, the inventors sampled with a cycle time of 73.52 picoseconds, and a requirement of acquiring time data from five cycles, and determined the phase in less than 0.5 microseconds (including processing time). This short cycle time can be particularly beneficial when measurements are being made, e.g., from a moving vehicle, such as in circumstances indicated in Provisional U.S. Patent Application No. 62/103,835. 
       FIG. 22  shows a flow diagram illustrating a method for determining a phase angle according to various embodiments of the disclosure based on the time-of-flight approach. This flow diagram is discussed with continuing reference to  FIGS. 1, 3, 4, 9, 12, 16, 20, and 21 . As shown, the process can include: 
     Process P 500 : obtaining reference signal  103  and RX signal  112 . 
     Process P 501 : passing both reference signal  103  and RX signal  112  through variable gain amplifier  141  ( FIG. 21 ) to generate an amplified reference A signal  145  and RX A signal  146 . 
     Process P 502 : pass both reference signal  103  and RX signal  112  through edge detector  142  ( FIG. 21 ) to convert these sinusoidal signals to square waves, ref. edge signal  147  and RX edge signal  148 . 
     Process P 503 : pass both ref. edge signal  147  and RX edge signal  148  to time-of-flight calculator  143  to determine a phase time  117 . 
     Process P 504 : evaluate phase time  117  at computing device (e.g., microprocessor  106 ) to determine quality of signal. 
     Process P 505 : if signal quality is poor (below threshold quality level), adjust gain on amplifier  141  to improve quality to edge detector  142 . 
     Process P 506 : if signal quality is good (above threshold quality level), record phase time  117  in computing device. 
     Process P 507 : compute phase angle between ref signal  103  and RX signal  112  and store at computing device for impedance calculation. 
       FIG. 23  depicts an illustrative environment including a sensor system  300  configured to secure complex impedance readings of an MUT  10 , and correlate the readings to a physical parameter of the MUT  10  according to various embodiments. To this extent, sensor system  200  includes a computing device  303  that can perform processes described herein in order to control measurement parameters and detect an impedance response of MUT  10 . In particular computing device  303  is shown as including a processing component  304 , and a storage component  305 , which makes sensor system  300  operable to detect an impedance response of MUT  10  and characterizing a physical parameter by performing any/all of the processes described herein and implementing any/all of the embodiments described herein. 
     Computing device  303  is shown including a processing component  304  (e.g., one or more processors), a storage component  305  (e.g., a storage hierarchy), an input/output pathway  306 , and an input/output (I/O) component  307 , which can connect to one or more I/O interfaces and/or devices such as input/output display  308 . In general, processing component  304  executes program code, which is at least partially fixed in storage component  305 . 
     While executing program code, processing component  304  can process data, which can result in reading and/or writing transformed data from/to the storage component  305  and/or the I/O component  307 , for further processing. The I/O Component  307  and/or I/O display  308 , can comprise one or more human I/O devices, which enable a human user(s)  309  to interact with the computing device  303 , and/or one or more communications devices to enable a system user(s) to communicate with computing device  303  using any type of communications link. To this extent, a Calibration Program  311 , the Measurement Program,  312 , the Impedance Program,  313 , and the MUT Characterization Program,  315 , can manage a set of interfaces (e.g., graphical user interface(s), application program interface, etc.) that enable human and/or system users to interact with the Computing Device and the Sensor System. 
     In any event, the Computing Device,  303 , can comprise one or more general purpose computing articles of manufacture (e.g., computing devices) capable of executing any of the stored program codes,  311 ,  312 ,  313 , and  314 , installed thereon. As used herein, it is understood that “program code” means any collection of instructions, in any language, code or notation, that cause a computing device having an information processing capability to perform a particular function either directly or after any combination of the following: (a) conversion to another language, code or notation; (b) reproduction in a different material form; and/or (c) decompression. 
     Further, the any of the stored programs can be implemented using a set of modules,  315 . In this case, a module,  315 , can enable the Computing Device,  303 , to perform a set of tasks used by any of the stored programs, and can be separately developed and/or implemented apart from other portions of impedance programs. As used herein, the term “component” means any configuration of hardware, with or without software, which implements the functionality described in conjunction therewith using any solution, while the term “module” means program code that enables the computer system Computing Device to implement the functionality described in conjunction therewith using any solution. When fixed in a Storage Component,  305 , of a Computing Device,  303 , that includes a Processing Component,  304 , a module is a substantial portion of a component that implements the functionality. Regardless, it is understood that two or more components, modules, and/or systems may share some/all of their respective hardware and/or software. Further, it is understood that some of the functionality discussed herein may not be implemented or additional functionality may be included as part of the Sensor System,  300 . 
     When the Sensor System  300  comprises multiple computing devices, each computing device may have only a portion of the stored programs fixed thereon (e.g., one or more modules  315 ). However, it is understood that the Sensor System,  300 , and stored processing programs are only representative of various possible equivalent computer systems that may perform a process described herein. To this extent, in other embodiments, the functionality provided by the Sensor System  300  and any of the stored processing programs,  311  through  315 , can be at least partially implemented by one or more computing devices that include any combination of general and/or specific purpose hardware with or without program code. In each embodiment, the hardware and program code, if included, can be created using standard engineering and programming techniques, respectively. 
     Regardless, when the Sensor System  300  includes multiple computing devices, the computing devices can communicate over any type of communications link. Further, while performing a process described herein, the Computing Device,  303 , of the Sensor System  300  can communicate with one or more other computing devices using any type of communications link. In either case, the communications link can comprise any combination of various types of wired and/or wireless links; comprise any combination of one or more types of networks; and/or utilize any combination of various types of transmission techniques and protocols. 
     The Sensor System  300  can obtain or provide data, such as Calibration Data  311  or Measured Levels and Phase Data  312 , for solution processing, e.g., by program(s)  312 ,  314 , or  315 . 
     While shown and described herein as a method and system for impedance detection and computation, it is understood that aspects of the invention further provide various alternative embodiments. For example, in one embodiment, the invention provides a computer program fixed in at least one computer-readable medium, which when executed, enables a computer system to control impedance detection and correlation parameters. To this extent, the computer readable medium includes program code, such as the impedance measurement program  312  ( FIG. 20 ), which implements some or all of the processes and/or embodiments described herein. It is understood that the term “computer readable medium” comprises one or more of any type of tangible medium of expression, now known or later developed, from which a copy of the program code can be perceived, reproduced, or otherwise communicated by a computing device. For example, the computer-readable medium can comprise: one or more portable storage articles of manufacture; one or more memory/storage components of a computing device; paper; etc.