Patent Publication Number: US-7590393-B2

Title: Low-noise transmitter system and method

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   This application is a continuation of U.S. patent application Ser. No. 10/927,311, filed Aug. 27, 2004, which claims the benefit of U.S. Provisional Patent Appl. No. 60/498,703, filed Aug. 29, 2003, each of which are incorporated by reference herein in their entireties. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to noise reduction in digital communication systems. 
   2. Related Art 
   In many communication systems, preventing interference between transmitted signals and received signals is essential. An example system is the Asymmetrical Digital Subscriber Line (ADSL) system. ADSL is a high-speed full duplex broadband transmission technique to connect a modem to the internet over ordinary telephone lines. For all ADSL applications, a hybrid circuit is used to couple receive (Rx) signals and transmit (Tx) signals from the telephone line. The hybrid circuit is also used to separate or decouple the Tx and Rx signals from each other. 
   Under ideal circumstances, the hybrid circuit is able to completely decouple the Tx and Rx signals from one another, so that the signal being transmitted does not interfere with the signal being received. However, due to imperfections in, for example, a telephone line and limited performance of the hybrid circuit, the hybrid circuit will not be able to completely decouple the Tx and Rx signals. Therefore, some amount of the residual Tx signal and Tx noise will be coupled onto the incoming Rx signal. 
   In order for the Rx signal to be correctly received, the interference from this Tx-to-Rx coupling must be minimized. This problem is exacerbated because the Rx signal is very small when compared to the Tx signal, and any residual Tx signal or noise can corrupt the Rx signal integrity. 
   As mentioned above, ADSL is only one type of communications system that experiences this problem. In many other communication systems, Tx noise needs to be minimized for similar reasons. In general, Tx noise must be limited so that the noise generated by a transmitter of a specific user does not interfere with the receiver of that same user (as in the case of ADSL), another user utilizing the same communications system, or another user using an entirely different communications system. 
   Although techniques exist for removing the residual Tx signal (e.g., echo cancellation), there are no comparable techniques to remove the residual Tx noise. Therefore, what is needed is a method and system for keeping Tx noise at a minimum. 
   SUMMARY OF THE INVENTION 
   A low-noise transmitter for use in a communications system such as ADSL includes a switched-current digital-to-analog converter (“DAC”) followed by a resistive transimpedance amplifier (“TIA”). The output of the DAC is connected to the low-impedance input of the TIA. As a result, there is no significant signal swing at the output of the DAC. At least one current source is coupled to the DAC to establish proper common-mode levels in the transmitter. Further, noise is reduced because there is no need to convert the DAC current into a voltage prior to feeding the signal to the resistive TIA. 
   In one embodiment, the current source is active and adds current to the transmitted signal without requiring conversion to a voltage prior to entering the TIA. In another embodiment, the current source is passive, such as a resistor connected to ground. Because the passive current source does not inject a signal into the system, use of a passive current source further reduces noise in the system compared to the active current source. Use of a passive current source also reduces required area and power consumption in the transmitter, while requiring no additional pins or external components. 
   In both embodiments, the value of the resistance of the current source is typically large enough such that the noise from an op-amp in the TIA is not significantly amplified at the output. In this manner, the overall output noise of the transmitter is reduced compared to previous transmitters. 
   Although the present invention will be described using the example of a transmitter, the present invention may also be used as an amplification system in any type of digital system. Further embodiments, features, and advantages of the present invention, as well as the structure and operation of the various embodiments of the present invention, are described in detail below with reference to the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES 
     The accompanying drawings, which are incorporated herein and form a part of the specification, illustrate the present invention and, together with the description, further serve to explain the principles of the invention and to enable a person skilled in the pertinent art to make and use the invention. 
       FIG. 1  is a diagram of an example transmitter that uses a switched-capacitor digital-to-analog converter (“DAC”) and a resistive gain amplifier (“RGA”). 
       FIG. 2  is a diagram of another example transmitter that uses an RGA. 
       FIG. 3  is a diagram of an example transmitter that uses a switched-current DAC and an RGA. 
       FIG. 4  is a diagram of an example transmitter that uses a switched-current DAC with an active current source and a resistive transimpedance amplifier (“TIA”). 
       FIG. 5  is a diagram of an example transmitter that uses a switched-current DAC with a passive current source and a resistive TIA. 
   

   The present invention will be described with reference to the accompanying drawings. The drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number. 
   DETAILED DESCRIPTION OF THE INVENTION  
   While specific configurations and arrangements are discussed, it should be understood that this is done for illustrative purposes only. A person skilled in the pertinent art will recognize that other configurations and arrangements can be used without departing from the spirit and scope of the present invention. It will be apparent to a person skilled in the pertinent art that this invention can also be employed in a variety of other applications. For example, one of ordinary skill in the relevant art will recognize that the amplification circuit described herein need not be used solely as an ADSL transmitter, but that it may be utilized as a low-noise amplifier in any application where an analog signal is amplified from either a digital or analog source. 
   When used as an ADSL transmitter, the specifications of the ADSL system require the transmitter to be able to drive a large signal with a linearity exceeding 90 dB. In order to meet this high-swing, high-linearity requirement, a digital-to-analog converter (“DAC”) followed by one or more stages of gain is used in various ADSL transmitter architectures. 
   Switched-capacitor DAC Followed by Resistive Gain Amplifier 
     FIG. 1  is a diagram of an example transmitter  100 . Transmitter  100  includes a switched-capacitor DAC (“SC-DAC”)  102  and a resistive gain amplifier (“RGA”)  104 . RGA  104  includes an op-amp  106 , resistors  108  and  110 , and feedback resistors  112  and  114 . Resistors  108  and  110  each have a resistance R 1 , while feedback resistors  112  and  114  each have a resistance R 2 . 
   The gain of RGA  104  is based on a ratio of feedback resistors  112  and  114  to resistors  108  and  110 . 
   SC-DAC  102  outputs a voltage-based signal V DAC . Op-amp  106  amplifies voltage-based signal V DAC , the gain set by the ratio of feedback resistors  112  and  114  and resistors  108  and  110 , and then outputs a voltage-based output signal V OUT . 
   SC-DACs, such as SC-DAC  102 , have relatively high noise levels due to the sampled-time nature of the switched capacitors. It is well known that switched-capacitor circuits have kT/C noise, which is noise due to fluctuation of charge stored on capacitance C. In order to reduce the power spectral density of this kT/C noise, either large capacitors must be used or very high oversampling ratios must be used. Large capacitors are impractical for integrated circuits because they require a large area. The oversampling ratio for an SC-DAC is limited by the speed of the circuit technology. Increasing the oversampling ratio of an SC-DAC usually requires a significant increase in power consumption. 
   Additionally, RGAs, such as RGA  104 , are typically power-consuming and noisy devices. Since SC-DAC  102  typically has relatively low output swing, RGA  104  must have a reasonable gain. As shown in the following equations, the noise at the output of RGA  104  contributed by op-amp  106  is amplified by RGA  104  in a similar manner to an input signal V DAC : 
   
     
       
         
           
             
               
                 
                   
                     
                       
                         V 
                         OUT 
                         2 
                       
                       _ 
                     
                     
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                       ⁢ 
                       
                           
                       
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                                 ⁢ 
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                                 ⁢ 
                                 
                                     
                                 
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                                   T 
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                                     R 
                                     1 
                                   
                                   · 
                                   
                                     A 
                                     RGA 
                                     2 
                                   
                                 
                               
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                                 ⁢ 
                                 
                                     
                                 
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                               + 
                               
                                 
                                   
                                     
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                                   ( 
                                   
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                   A 
                   RGA 
                 
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                       OUT 
                     
                     
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                       DAC 
                     
                   
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                         R 
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                 ( 
                 
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   A RGA  is the amplitude of the amplifier noise, and is a function of the gain of the amplifier. V N  is the voltage input into op-amp  106 . k is Boltzmann&#39;s constant, T is the temperature in ° K, and C is the capacitance of the SC-DAC. Each of the resistor sets (i.e., resistors  108  and  110 , and feedback resistors  112  and  114 ) adds noise to the system, as shown by the inclusion of R 1  and R 2  in Eq. 1. This puts a stringent noise specification upon op-amp  106  within RGA  104 . 
     FIG. 2  is a diagram of an example transmitter  200 . Transmitter  200  is similar in architecture to transmitter  100 , but uses a generic low-impedance input source  202  in place of an SC-DAC. Transmitter  200  also includes an RGA  204  having an op-amp  206 , resistors  208  and  210 , feedback resistors  212  and  214 , and voltage sources  216  and  218 . Resistors  208  and  210  are approximately equal, each having a resistance R 1 . Feedback resistors  212  and  214  are also approximately equal to each other, each having a resistance R 2 . 
   Op-amp  206  outputs an amplified signal VOUT, wherein the gain is based on the ratio of feedback resistors  212  and  214  to resistors  208  and  210 . 
   Switched-current DAC with Resistive Load Followed by RGA 
     FIG. 3  is a diagram of an example transmitter  300  that uses a switched-current DAC (“SI-DAC”)  302  with an RGA  304  and a resistive load created by resistors  316  and  318 . Like RGA  104  and RGA  204 , RGA  304  includes an op-amp  306 , resistors  308  and  310 , and feedback resistors  312  and  314 . Since an SI-DAC is used in place of an SC-DAC, resistors  316  and  318  are used to set the voltage swing at the output of the SI-DAC. Compared to the example of  FIGS. 1 and 2 , SI-DAC  302  followed by RGA  304  has a lower noise. This is because SI-DAC  302  does not add the kT/C noise of an SC-DAC. Even so, this architecture has several limitations. 
   First, transmitter  300  utilizes an RGA. Therefore, all the problems described with respect to RGA  104  and RGA  204  apply to RGA  304 . 
   Specifically, the signal is still affected by two sets of resistors (i.e., resistors  308  and  310 , and feedback resistors  312  and  314 ) to create an output signal, which adds noise to the system. 
   Also, a limitation with high-linearity SI-DACs is that their output swing is typically severely limited. As shown in Eq. 1, if V OUT  is assumed to be fixed, requirements are imposed on the product of V DAC  and A RGA . A large A RGA  increases the total output noise of the transmitter. Therefore, transmitter  300  requires a reasonably large V DAC . This can be difficult to achieve while maintaining high linearity. 
   In each of transmitters  100 ,  200 , and  300 , the act of converting the signal between current and voltage domains results in a large amount of output noise due to the resistors. Also, resistance R 1  from the resistors presents a load that needs to be driven by the previous stage. Thus, if the resistors could be removed, the noise and total power consumption of the overall circuit would be significantly reduced. 
   Switched-current DAC with Active Current Source 
     FIG. 4  is a diagram of an example transmitter  400  according to an embodiment of the present invention. Transmitter  400  includes an SI-DAC  402 , active current sources  404  and  406 , and a resistive transimpedance amplifier (“TIA”)  408 . TIA  408  includes an op-amp  410 , feedback resistors  412  and  414 , and voltage sources  416  and  418 . SI-DAC  402  converts an input digital signal to an analog current-based signal. Because a TIA can convert a current-based signal to a voltage-based signal within the TIA itself, there is no need for resistors to perform this step before the signal is input to the TIA. The current-based signal from SI-DAC  402  can be input directly into resistive TIA  408 . 
   Thus, an SI-DAC with active current sources  404  and  406  followed by a resistive TIA avoids many of the problems described above. For example, an input signal in transmitter  400  is affected by one less resistor set. This removes any noise associated with resistors in the examples of  FIGS. 1 ,  2 , and  3 . Instead of amplifying an input voltage, resistive TIA  408  acts as a current-to-voltage converter and converts the current-based signal from SI-DAC  402  into a voltage-based signal. Resistive TIA  408  also amplifies the signal to produce an output signal V OUT . Thus, as shown in the following noise equation, the output-referred amplifier noise is no longer a function of the gain of the amplifier stage: 
                       V   OUT   2     _       Δ   ⁢           ⁢   f       =     2   ·       (       4   ⁢   k   ⁢           ⁢     T   ·     R   TIA         +         V   N   2     _       Δ   ⁢           ⁢   f       +           I   CS   2     _       Δ   ⁢           ⁢   f       ·     R   TIA   2       +           I   DAC   2     _       Δ   ⁢           ⁢   f       ·     R   TIA   2         )     .               (     Eq   .           ⁢   3     )               
Further, the transimpedance gain is equal to the resistance of feedback resistors  412  and  414 , or R TIA , rather than a ratio of feedback resistors to resistors.
 
   The inputs to the TIA create a virtual ground node at node  420 . Since SI-DAC  402  is feeding into a virtual ground node, the signal swing at the output of SI-DAC  402  is kept small, even if a large output swing exists. Therefore, high linearity of the transmitter output V OUT  can be maintained. Current sources  404  and  406  shown in  FIG. 4  maintain proper common-mode levels throughout the transmitter and drive the analog current-based signal. 
   Current sources  404  and  406  shown in the embodiment of  FIG. 4  are created from active devices. Because the active devices inject a signal into a node  420  between SI-DAC  402  and resistive TIA  408 , they may contribute noise to the output of transmitter  400 . In addition, the active devices may be biased by a network of similar active devices. These bias devices could contribute additional noise to the output. The noise produced by these devices can cause difficulties at low frequencies, where the flicker noise of the active devices can dominate, since the I cs   2 /Δf term becomes very large. Since the ADSL transmit band, for example, is at approximately 100 kHz, the flicker noise of the active devices may overpower the Tx signal in such an application. 
   In order to limit the total noise contributed by the active devices, several steps can be taken. Flicker noise can be reduced by increasing the area of the active devices, and/or the noise from the bias devices can be effectively eliminated by attaching a large bypass capacitor to the bias network. However, increased area translates into increased parasitic capacitances. Further, attaching a bypass capacitor would require an additional pin on the package and an additional discrete component. Both of these steps would incur additional system costs. 
   Switched-current DAC with Passive Current Source 
     FIG. 5  is a diagram of an example transmitter  500  according to another embodiment of the present invention. Transmitter  500  includes an SI-DAC  502  with passive current sources  504  and  506  followed by a resistive TIA  508 . TIA  508  includes an op-amp  510 , feedback resistors  512  and  514 , and voltage sources  516  and  518 . Passive current sources  504  and  506  may be, for example, resistors connected to ground which are used to establish proper common-mode levels. In the example of  FIG. 5 , each of passive current sources  504  and  506  has a resistance R CS . 
   Compared to transmitter  400 , transmitter  500  has reduced noise due to the substitution of passive current sources for active current sources. Like transmitter  400 , the signal in transmitter  500  is processed by one less resistor set compared to transmitters  100 ,  200 , and  300 , due to the lack of resistors. As shown in  FIG. 5 , current sources  504  and  506  branch off of a signal path  520  between SI-DAC  502  and TIA  508 . Although a portion of the signal output by SI-DAC  502  may enter passive current sources  504  and  506 , that portion of the signal is common-mode and hence not processed by TIA  508 . Therefore, the differential transfer function of the signal is unaffected by the presence of passive current sources  504  and  506 . 
   Thus, the total output noise of the embodiment of  FIG. 5  is: 
   
     
       
         
           
             
               
                 
                   
                     
                       
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                         OUT 
                         2 
                       
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                       ⁢ 
                       
                           
                       
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                       ) 
                     
                   
                 
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                 where 
               
             
             
               
                 ( 
                 
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                     TIA 
                   
                   
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                     CS 
                   
                 
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                 1. 
               
             
             
               
                 ( 
                 
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                   5 
                 
                 ) 
               
             
           
         
       
     
   
   The value of resistance Rcs used in passive current sources  504  and  506  is typically large enough such that noise from op-amp  510  in TIA  508  is not significantly amplified at the output. For ADSL applications, R CS  may be, for example, 1 kΩ. Also, passive current sources  504  and  506  have negligible flicker noise. Therefore, at low frequencies, the overall output noise of this topology is typically far lower than in systems described above. This allows improved application at lower frequencies than with previous systems. For instance, for ADSL applications the frequency range may be, for example, 20 kHz to 276 kHz. However, one of ordinary skill in the relevant art(s) will recognize that op-amp  510  may be tuned to any frequency for the same or alternate applications without departing from the spirit and scope of the present invention. Also, passive current sources  504  are much smaller in area and power consumption than active current sources, and require no additional pins or external components. Therefore, the embodiment of  FIG. 5  is amenable to a low-cost, low-noise system, and effectively reduces noise from a Tx signal. 
   The embodiments described with reference to  FIGS. 4 and 5  are also different from that of transmitter  300  in that the performance of the embodiments is quite different. First, transmitters  400  and  500  inject the signal from the SI-DAC into the TIA virtual ground formed by the current sources, so there is no significant signal swing at the output of the SI-DAC. Additionally, a resistive TIA amplifies op-amp noise far less than, for example, RGA  304  in transmitter  300 . In transmitter  500 , the purpose of resistors R cs  used as passive current sources  504  and  506  is not to convert the current from SI-DAC  502  into a voltage; instead, resistors R cs , are used as passive current sources to establish proper common mode. In such ways, the Tx noise in each of example transmitters  400  and  500  is reduced compared to example transmitters  100 ,  200 , and  300 . 
   Conclusion 
   While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail can be made therein without departing from the spirit and scope of the invention. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.