Patent Publication Number: US-7221229-B2

Title: Receiver circuit having an optical reception device

Description:
RELATED APPLICATION 
     The present application claims priority of U.S. Patent Application Ser. No. 60/540,870 filed by Karl Schrodinger on Jan. 30, 2004. 
    
    
     FIELD OF THE INVENTION 
     The invention relates to a receiver circuit having an optical reception device and having an amplifier connected downstream of the optical reception device. In particular, the invention relates to a receiver circuit having a transimpedance amplifier for optical transmission systems. 
     BACKGROUND OF THE INVENTION 
     Receiver circuit having an optical reception device are known in which light incident on the optical reception device—for example, light from an optical waveguide of an optical data transmission system—is detected by the optical reception device with formation of an electrical signal (e.g. a photocurrent) and the electrical signal is subsequently amplified by the amplifier connected downstream. 
     An optical receiver circuit having an optical reception device and having an amplifier connected downstream is described for example in the article “High Gain Transimpedance Amplifier in InP-Based HBT Technology for the Receiver in 40-Gb/s Optical-Fiber TDM Links” (Jens Müllrich, Herbert Thurner, Ernst Müllner, Joseph F. Jensen, Senior Member, IEEE, William E. Stanchina, Member, IEEE, M. Kardos, and Hans-Martin Rein, Senior Member, IEEE—IEEE Journal of Solid State Circuits, vol. 35, No. 9, September 2000, pages 1260 to 1265). In the case of this receiver circuit, at the input end there is a differentially operated transimpedance amplifier—that is to say a differential amplifier—connected by one input to a photodiode as reception device. The other input of the differentially operated transimpedance amplifier is connected to a DC amplifier which feeds a “correction current” into the differential amplifier for the purpose of offset correction of the photocurrent of the photodiode. The magnitude of this “correction current” that is fed in amounts to half the current swing of the photodiode during operation. 
     An optical receiver circuit is always subject to noise. In the case of an optical receiver circuit having a transimpedance amplifier, the most important noise sources are the input transistor of the transimpedance amplifier and the transimpedance impedance. 
     There is a need for receiver circuits which have a favorable noise behavior. 
     SUMMARY OF THE INVENTION 
     The invention provides a receiver circuit, which has: an optical reception device and an amplifier connected to the reception device, the amplifier having a circuit for setting the operating point of the amplifier and also at least one control terminal of the circuit, by means of which the operating point of the amplifier can be changed over between at least two values at the user end (i.e., the operating point is user configurable). 
     The present invention is based on the concept of providing an amplifier operating point that can be selectively changed over for the purpose of noise optimization. In this case, it is preferably the operating point of an input transistor of the amplifier that is set, the noise of which predominates over the noise of the amplifier. In this case, the noise of the input transistor can be set by way of the operating point thereof. 
     In a preferred refinement, the circuit for setting the operating point of the amplifier forms a setting of the operating point of the input transistor by setting the current in the input transistor. In this case, the circuit for setting the operating point of the amplifier is preferably formed between the input transistor and a reference point, at which the operating voltage is present. Changing over the current in the input transistor changes over the operating point thereof. This is accompanied by an altered noise, in which case it holds true that the noise in the input transistor likewise decreases as the current in the transistor decreases. 
     Preferably, the circuit for setting the operating point of the amplifier is formed by an impedance network with at least one switching device, which can be changed over at the user end by means of the at least one control terminal, the total impedance of the impedance network being altered. In particular, it is preferably provided that the impedance network has a plurality of ohmic resistors, which can be connected in and disconnected by means of the at least one switching device and the at least one control terminal. 
     In one development of the invention, the amplifier furthermore has at least one gain control terminal, by means of which the gain of the amplifier can be changed over at least between two gain values at the user end. This enables an optimal optical sensitivity: this is because the adjustability of the gain of the amplifier makes it possible to set the maximum gain of the amplifier depending on the prescribed bandwidth, or bandwidth to be achieved, of the receiver circuit. By way of example, on account of the approximately constant bandwidth (B)-gain (V) product (B*V=K; K results from the individual configuration of the receiver circuit), it is possible to set the maximum gain V and thus the maximum sensitivity of the receiver circuit by choosing V=K/B. The receiver circuit can thus be used optimally for different data rates. Thus, on account of the gain that can be changed over, the receiver circuit can be individually adapted for example to transmission rates of 1 Gbps (gigabit per second), 2 Gbps or 4 Gbps. 
     A further essential advantage of the receiver circuit with a gain that can be changed over consists in its optimal noise behavior. By way of example, if a photodiode is used as the reception device and a transimpedance amplifier is used as the amplifier, then the current noise has a particularly relevant part to play in the amplifier. However, the current noise which is attributable to the transimpedance amplifier generally becomes lower toward higher gains of the amplifier, so that, when the optimum—that is to say maximum—gain is chosen, the current noise of the amplifier also decreases. However, with other types of amplifier, too, it generally holds true that the signal-to-noise ratio becomes better in the case of a higher gain. In summary, the noise behavior of the receiver circuit can be improved further as a result of the user-end setting of the optimum gain value depending on the respective bandwidth requirement. 
     The amplifier preferably has a feedback impedance, which influences the gain of the amplifier. The impedance of the feedback impedance can then be set externally at the user end by means of the at least one gain control terminal. In particular, the resistance of the feedback impedance should be able to be set at the user end by means of the at least one control terminal. 
     In order to be able to ensure the adjustability of the impedance of the feedback impedance in a particularly simple manner, one advantageous development of the receiver circuit proposes that the feedback impedance is formed by an impedance network with at least one switching device, which can be changed over at the user end by means of the at least one control terminal and which alters the impedance or the resistance of the impedance network in the case of a changeover. In this case, the switching device is preferably formed by a switching transistor, in particular a MOS-FET transistor. 
     Another advantageous development of the receiver circuit proposes that the feedback impedance is formed by an impedance network with at least one variable impedance, the impedance of which can be set at the user end within a predetermined impedance range at least approximately linearly by means of the control terminal. The variable impedance may be formed for example by a transistor, in particular a MOS-FET transistor. 
     In transimpedance amplifiers, the bandwidth is approximately proportional to the reciprocal of the feedback impedance, that is to say to 1/feedback impedance, since the gain is proportional to the feedback impedance. In this case, the gain is determined by the so-called transimpedance (=output voltage/input current). 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention is explained in more detail below using an exemplary embodiment with reference to the figures, in which: 
         FIG. 1  shows an exemplary embodiment of a receiver circuit having an optical reception device and an amplifier connected to the reception device, said amplifier having a feedback impedance; 
         FIG. 2  shows an exemplary embodiment of the feedback impedance of the amplifier of the receiver circuit of  FIG. 1 ; 
         FIG. 3  shows an exemplary embodiment of the amplifier of the receiver circuit of  FIG. 1 ; and 
         FIG. 4  shows the spectral noise power of the amplifier of  FIG. 3  for three different circuit states of the amplifier. 
     
    
    
     DESCRIPTION OF A PREFERRED EXEMPLARY EMBODIMENT 
       FIG. 1  reveals a receiver circuit  10  with a photodiode  20  as optical reception device. A transimpedance amplifier  30  is arranged downstream of the photodiode  20 . The transimpedance amplifier  30  comprises a voltage amplifier  40 , for example an operational amplifier, and a feedback impedance  50 . The feedback impedance  50  is connected to the input end of the operational amplifier  40  by its terminal E 50  and to the output end of the operational amplifier  40  by its terminal A 50 . 
     At the output end, the transimpedance amplifier  30  is additionally connected to a differential amplifier  60 , which amplifies the output signal Sa of the transimpedance amplifier  30 . A further amplification of the signal is effected by a second differential amplifier  70  arranged downstream of the first differential amplifier  60 . 
       FIG. 1  furthermore reveals a control circuit  80 , which, at the input end, is connected to the two outputs A 70   a  and A 70   b  of the second differential amplifier  70 . The control circuit  80  additionally has a control input S 80 , via which a user-end control signal Sb can be fed into the control circuit  80 . The control input S 80  thus forms a control terminal S 10  of the receiver circuit  10 . 
     By an output A 80 , the control circuit  80  is connected to a control terminal S 30  of the transimpedance amplifier  30  and thus to a control input S 50  of the feedback impedance  50 . Via said control input S 50 , the control circuit  80  can define the impedance, in particular also the resistance, of the feedback impedance  50  by means of an impedance specification signal Sr formed from the user-end control signal Sb. 
     Furthermore, the optical receiver circuit is equipped with a DCC circuit  90  (DCC: Duty Cycle Control), which effects a control of the optical receiver circuit. The DCC circuit  90  or the duty cycle control (offset control) formed by it controls the sampling threshold for the downstream differential amplifiers, so that the signal is sampled at the 50% value of the amplitude and, as a result, no signal pulse distortions (duty cycle) are produced. This can be effected by feeding a current into a respective one of the preamplifiers (transimpedance amplifiers) or else by feeding in a voltage at the inputs of the differential amplifiers directly. 
     The photodiode  20  is connected via a low-pass filter  100  formed from a capacitor C PD  and a resistor R PD , a supply voltage VCC 1  being applied to said filter. The low-pass filter  100  serves to “filter out” possible interference signals on the supply voltage VCC. 
     The optical receiver circuit  10  in accordance with  FIG. 1  is operated as follows: 
     When light is incident, a photocurrent I photo  is generated by the photodiode  20  and fed into the transimpedance amplifier  30 , where the photocurrent is amplified to form the output signal Sa. The electrical output signal Sa is amplified further by the two differential amplifiers  60  and  70  to form an amplified output signal Sa′ and passes to the output A 10  of the optical receiver circuit  10 ; the output A 10  of the optical receiver circuit  10  is thus formed by the two outputs A 70   a  and A 70   b  of the second differential amplifier  70 . 
     The gain of the transimpedance amplifier  30  is set at the user end by means of the control signal Sb via the control terminal S 80  of the control circuit  80  or via the control terminal S 10  of the receiver circuit  10 . For this purpose, the control signal Sb generated at the user end passes to the control circuit  80 , which, with its impedance specification signal Sr, sets the resistance of the feedback impedance  50 . This is because the magnitude of the resistance (|R|) of the feedback impedance  50  directly influences the gain of the transimpedance amplifier  30  because the following holds true:
 
 Sa=|R|*I   photo 
 
     Thus, in the case of the arrangement in accordance with  FIG. 1 , the gain of the transimpedance amplifier  30  can be prescribed at the user end by means of the control signal Sb. 
     When prescribing an optimum gain value for the transimpedance amplifier  30 , it is necessary to take account of the bandwidth B respectively required. In concrete terms, a very large gain is possible given a very small bandwidth, whereas only a very small gain can be achieved given a very large bandwidth. In concrete terms, this is due to the fact that, to a first approximation, the bandwidth-gain product (V*B) of the receiver circuit  10  is approximately constant and is prescribed by the individual configuration of the receiver circuit. The product V*B can be determined by measurement, for example. 
     Thus, if a specific bandwidth is prescribed or is at least to be achieved, then the maximum permissible gain can be derived from this at the user end. A corresponding gain value is then set by the control circuit  80  through the selection of the corresponding magnitude of the feedback impedance  50 . 
     The desired gain can therefore be prescribed at the user end via the control input S 80  and thus by means of the control signal Sb. As an alternative—given a corresponding configuration of the control circuit  80 —a bandwidth to be achieved can also be communicated to the control circuit  80  at the user end by means of the control signal Sb, from which the maximum permissible gain V is then determined by the control circuit  80  in accordance with the mathematical relationship mentioned above and is communicated to the transimpedance amplifier  30  via the output A 80  and the control terminal S 50 . 
     In connection with  FIG. 1 , the user-end control signal Sb was conducted to the transimpedance amplifier  30  via the control device  80 . Instead of this, the user-end control signal Sb may also be applied directly to the control terminal S 30  of the transimpedance amplifier  30 . 
     Moreover, the transimpedance amplifier  30 , the two differential amplifiers  60  and  70 , the control circuit  80  and the DCC circuit  90  may also be regarded as one “amplifier unit” or as one “amplifier” whose control terminal for feeding in the user-end control signal Sb is formed by the terminal S 80  of the control circuit  80 . 
       FIG. 2  illustrates an exemplary embodiment of a feedback impedance  50  in accordance with  FIG. 1 . The feedback impedance is formed by user configurable impedance network. The illustration reveals an ohmic resistor RF 1 , with which capacitors CF 1 , CF 2 , CF 3 , CFC 1  and CFC 2  are connected in parallel. In addition, further ohmic resistors RF 2  and RF 3  are connected in parallel with the resistor RF 1 . 
     As can be discerned in  FIG. 2 , the resistor RF 2  and the capacitor CF 2  are connected in parallel and are connected to a switching transistor  210 . If the switching transistor  210  is switched off, then the resistor RF 2  and the capacitor CF 2  play no part in the total impedance of the impedance network. By contrast, if the switching transistor  210  is switched on, then the resistors RF 1  and RF 2  form an ohmic parallel connection, with the result that the total resistance of the impedance network is reduced. The capacitor CF 2  correspondingly increases the total capacitance of the impedance network since the capacitor CF 2  is added to the capacitor CF 1 . 
     The resistor RF 3  and the capacitor CF 3  can be connected in parallel with the first resistor RF 1  in a corresponding manner by means of a second switching transistor  220 . 
       FIG. 2  furthermore reveals a MOS-FET transistor  230 , which represents a linearly controllable resistor. Depending on the gate voltage applied to the MOS-FET transistor, a transistor resistor is produced which is connected in parallel with the first resistor RF 1  and thus linearly reduces the total resistance of the impedance network. The resistance of the impedance network can be set in a continuously variable manner by application of the gate voltage. 
     Via a third switching transistor  240  and a fourth switching transistor  250 , the capacitor CFC 1  and the capacitor CFC 2  can likewise be connected in parallel with the first resistor RF 1 , or else “disconnected”. 
       FIG. 2  furthermore reveals a coding device  300 , the input E 300  of which forms the control terminal S 50  of the feedback impedance  50  in accordance with  FIG. 1 . At the output end, the coding device  300  is connected to the four switching transistors  210 ,  220 ,  240  and  250  and also to the linearly operating MOS-FET transistor  230 . 
     The coding device  300  serves to recode the impedance specification signal Sr formed by the control circuit  80  in such a way that the feedback impedance  50  or the impedance network forms the desired impedance and the transimpedance amplifier  30  thus achieves the required gain. 
     The impedance network is driven as follows for the operation of the receiver circuit in accordance with  FIG. 1 : 
     The resistor RF 1  serves for setting the largest gain and thus the smallest bandwidth of the transimpedance amplifier  30 . In this operating mode—that is to say with the smallest bandwidth—the second resistor RF 2  and the third resistor RF 3  are disconnected by the two switching transistors  210  and  220 . The capacitor CF 1  serves for compensation against oscillation tendencies of the receiver circuit  10 . 
     If a higher data rate is required, then the second resistor RF 2  is connected in, by way of example; a lower transimpedance impedance is thus produced as a result of the two resistors RF 1  and RF 2  being connected in parallel, as a result of which the gain of the transimpedance amplifier  30  is reduced and the bandwidth is increased. 
     As a result of further connection—for example of the third resistor RF 3 —the resistance of the feedback impedance  50  and thus the gain of the transimpedance amplifier  30  can be reduced further, as a result of which the bandwidth is increased further. The compensation capacitors CF 2  and CF 3  that are necessary, if appropriate, for compensation against oscillation tendencies are additionally connected in at the same time as the two resistors RF 2  and RF 3  by the two switching transistors  210  and  220 . In this case, the transistors  210 ,  220 ,  230 ,  240  and  250  are changed over by the control signal SV by means of the coding device  300 . 
     The function of the MOS-FET transistor  230 , which is likewise controlled by the coding device  300  and the control circuit  80 , serves primarily for amplitude control. If the output power of the transimpedance amplifier rises increasingly, then the transistor  230  is driven linearly, so that the feedback impedance (transimpedance impedance)  50  of the transimpedance amplifier  30  is continuously decreased: overdriving of the transimpedance amplifier  30  can be prevented in this way. In order to be able to identify an increase in the output power of the transimpedance amplifier  30 , the control circuit  80  in accordance with  FIG. 1  is connected to the output signals Sa′ and −Sa′ of the further differential amplifier  70 . 
     The additional capacitors CFC 1  and CFC 2  can be connected in with the associated switching transistors  240  and  250  in order to avoid oscillations; this may be necessary particularly when the feedback impedance  50  of the transimpedance amplifier  30  is decreased linearly on account of the MOS-FET transistor  230 . 
     In summary, in the case of the exemplary embodiment in accordance with  FIG. 2 , the feedback impedance  50  is reduced by resistors and/or capacitors being connected in “parallel”. Instead of this or in addition, a changeover of the impedance of the feedback impedance  50  may also be achieved through a series circuit of connectable resistors and/or connectable capacitors. 
     The coding device  300  may be formed for example by an integrated circuit which correspondingly converts the impedance specification signal Sr in such a way that the transistors  210 ,  220 ,  230 ,  240  and  250  are driven in the manner explained above. 
       FIG. 3  shows in detail an exemplary embodiment of the transimpedance amplifier  30  illustrated in  FIG. 1 . In this case, the feedback impedance  30  may be formed by an impedance network in a manner corresponding to  FIG. 2 . In the exemplary embodiment of  FIG. 3 , the output signal Sa of the transimpedance amplifier  30  is present at the reference point  300  of the circuit. 
     The photodiode  20  serving to detect optical signals is assigned a capacitance C IN , which comprises the input capacitance of the photodiode  20  and also parasitic capacitances. The capacitance C IN , together with the feedback impedance RF  50 , forms a low-pass filter whose limiting frequency is determined by the equation f=1/(2πC IN *RF). As the impedance RF increases, the bandwidth of the amplifier is thus reduced, i.e. the maximum data rate which the amplifier  30  can amplify decreases as the value of the impedance RF increases. 
     The transistors of the amplifier  30  which are explained below are embodied using bipolar technology. However, they may also be embodied as field-effect transistors in a corresponding manner. 
     A first transistor T 1  is present, the base (control) terminal of which is connected to the photodiode  20 . The emitter (first) terminal of the transistor T 1  is connected to ground. The collector (second) terminal of the transistor T 1  is connected to the operating voltage U B , present at a reference point  130 , via a plurality of resistors R C1 , R C2 , R C3 . The resistors R C1 , R C2 , and R C3  are connected in parallel with one another and together form resistance R C . Two of the resistors R C2  R C3  are in this case formed such that they can be connected in or disconnected by switches M 1 , M 2  via corresponding control terminals S 1 , S 2 . In this case, the switches M 1 , M 2  are formed as MOS transistors, the control terminals S 1 , S 2  being connected to the respective gate terminal. However, they may also be embodied differently. 
     Instead of or in addition to the resistors R C1 , R C2 , R C3  being connected in parallel in the manner illustrated, it is possible to achieve a resistor setting of the resistor R C  at the collector terminal also by means of a series circuit of connectable resistors and/or connectable capacitors. 
     The collector terminal of the transistor T 1  is furthermore connected to the base (control) terminal of a second transistor T 2 . The collector (second) terminal thereof is directly connected to the operating voltage. The emitter (first) terminal of the transistor T 2  is connected to ground via a mirror circuit  120  having two further transistor T 3 , T 4 . The mirror circuit  120  serves for setting a current for the transistor T 2 . This function may also be provided by other components, for example an ohmic resistor. 
     The emitter terminal of the transistor T 2  is connected to the base terminal of the transistor T 1  via the feedback impedance RF. An output signal Sa (cf.  FIG. 1 ) of the amplifier  30  is tapped off at the node  300 . 
     The circuit of  FIG. 3  functions as follows: 
     The base-emitter voltage U BE1  of the first transistor T 1  and the base-emitter voltage U BE2  of the second transistor T 2  are approximately constant. This results from the feedback to the base terminal of the first transistors T 1  via the impedance RF. Since the voltage drop U RC  across the resistor R C  (comprising one or a plurality of the parallel-connected resistors R C1 , R C2 , R C3 ) depends only on the constant operating voltage U B  and the two base-emitter voltages U BE1 , U BE2  of T 1  and T 2 , which are likewise constant to a first approximation (U RC =U B −U BE1 −U BE2 ), the resistor R C  alone determines the current in the transistor T 1 . In the event of supplementarily connecting or disconnecting resistors R C2 , R C3  via the switches M 1 , M 2 , it is thus possible to set the current through the transistor T 1 . 
     As a result, the operating point of the input transistor T 1  is also set insofar as the latter acquires, depending of R C , a value between maximum voltage, (operating voltage U B ) and minimum voltage (ground). In this case, the term operating point denotes the quiescent state in the absence of an input signal. This is described by a specific point on the characteristic curve of the transistor. 
     The input signal of the photodiode  20  is amplified first by the transistor T 1  and then by the transistor T 2 . The current through the transistor T 2  is in this case set by the mirror circuit  120 , on which a reference current I B  is impressed. 
     Consideration shall now be given firstly to the case where the received data have a high data rate or bandwidth. A high bandwidth is accompanied by a low gain and a low value of RF. If a lower data rate is then present, the bandwidth of the amplifier may decrease. For this purpose, it is possible, on the one hand, for the value of RF to be chosen to be larger (cf  FIG. 2 ), which leads to a larger gain and a smaller bandwidth. Furthermore, a smaller bandwidth of the amplifier may be achieved by the resistor R C  between the collector terminal of the transistor T 1  and the operating voltage U B  being chosen to have a higher value. This is done by disconnecting one or a plurality of the parallel-connected resistors R C2 , R C3  by means of corresponding control signals on the control terminals S 1 , S 2 . 
     The situation, then, is such that the noise of the amplifier  30  is determined on the one hand by the noise of the transimpedance impedance RF and on the other hand by the noise of the transistor T 1  and the operating point thereof. In this case, it holds true that as the current through the transistor T 1  increases, the noise also increases. These relationships are described for example in H. Kressel (Editor), Semiconductor Devices for Optical Communication, 2 nd  Edition, Springer Verlag 1982, page 89 et seq. 
     Disconnecting one or a plurality of the resistors R C2 , R C3  increases the total resistance R C  between the collector terminal of the transistor T 1  and the operating voltage. This increased total resistance R C  leads to a lower current through the transistor T 1  and an altered operating point of the transistor T 1 . Since the noise of the transistor T 1  decreases as the current through the transistor T 1  decreases, the noise of the amplifier  30  can be reduced in this way. 
     These relationships are explained in greater detail with reference to  FIG. 4 . In  FIG. 4 , the “0” curve  200  specifies the original noise curve. The illustration shows the spectral noise power as a function of frequency. Curve  200  reveals that essentially two components make a contribution to the noise of the amplifier  30 . On the one hand, at low frequencies in the region A, a contribution to the noise is made by the noise level of the feedback impedance R F . The noise of the amplifier predominates at high frequencies in the region B. In between is a transition region C, in which the noise increases towards higher frequencies. 
     In the event of a changeover of the gain of the amplifier  30  as described in  FIGS. 1 and 2 , the noise level of the transimpedance impedance RF and thus the low-frequency noise component are reduced. In concrete terms, the value of the impedance RF is switched higher for the case of lower data rates. The gain of the amplifier  30  is increased in this case. At the same time, the bandwidth of the amplifier  30  is reduced, said bandwidth being approximately inversely proportional to the gain. This is connected with the fact that, at an increased value of RF, the limiting frequency of the low-pass filter comprising the capacitance C IN  and RF decreases. Furthermore, an increase in the transimpedance impedance RF leads to a reduced noise influence of the impedance RF. Thus, the noise &lt;I R   2 &gt; of the impedance in the transimpedance amplifier is defined as &lt;I R   2 &gt;=(4 kT/FR)Δf, where k is equal to Boltzmann&#39;s constant and T specifies the temperature. At a constant temperature, the noise influence of the impedance thus decreases as the impedance increases. 
     This effect is illustrated in  FIG. 4 . The “1” curve  310 , which specifies the spectral noise power after a changeover of the transimpedance impedance RF to a higher impedance, is lowered in the region of low frequencies. 
     A further reduction of the noise is exhibited by the “2” curve  320 , which specifies the spectral noise power after the changeover of the current in the transistor T 1  through disconnection of one or a plurality of the resistors R C2 , R C3 , i.e., after increasing the resistance R C  in the collector arm. Increasing the resistance R C  reduces the current in the transistor T 1 . This reduction of the current in the transistor T 1  leads to a reduced noise power of the transistor T 1 , which is manifested in a reduced noise power of the curve  320  at high frequencies, wherein the noise power of the transistor predominates over the noise of the amplifier. At the same time, the bandwidth of the amplifier is also reduced in the case of a resistance R C  having a higher value. This effect also takes place, as explained, when the transimpedance impedance RF is increased. 
     Consequently, an amplifier circuit  30  is described which provides a noise optimization of the amplifier on the one hand by means of a changeover of the gain of the amplifier  30  (and thus a changeover of the bandwidth of the amplifier  30 ) and on the other hand by means of an operating point changeover in the input transistor T 1  of the amplifier  30  at lower data rates. 
     The configuration of the invention is not restricted to the exemplary embodiments present above, which are to be understood merely by way of example. The person skilled in the art recognizes that numerous alternative embodiment variants exist which, despite their deviation from the exemplary embodiments described, make use of the teaching defined in the claims below.