Patent Publication Number: US-10763990-B1

Title: Switching frequency methods and apparatus for ambient backscatter networking and jamming

Description:
BACKGROUND OF THE INVENTION 
     The present invention is in the technical field of wireless transmission and reception of information. More particularly, the present invention is in the technical field of ultra-low power backscatter radio communication. Embodiments and aspects of part of the invention described herein, focus on digital bistatic backscatter radio; the illuminator of a tag and the receiver of the backscattered information (from the tag) are distinct units, located at different points in space. 
     The illuminating signal may be modulated and/or emitted from legacy wireless systems or infrastructure (ambient case). 
     Embodiments and aspects of part of the invention described herein, are inspired by recent findings in digital bistatic backscatter radio [Bletsas et al 12a], [Bletsas et al 12b], [Bletsas et al 13] (where the illuminating signal is unmodulated) and ambient backscatter radio (where the illuminating signal is modulated and carries its own information). Although easy and straightforward to implement, current methods [Smith et al 13] and [Smith et al 14] either exploit on-off keying (OOK) modulation at the tag or utilize detection schemes with significant trade-offs between communication range and information transmission rate. Additionally, the structure and/or modulation of the ambient signal is explicitly considered. Moreover, current methods require multiple access from multiple tags utilizing time division multiple access (TDMA) or code division multiple access (CDMA), which increases complexity and cost of the overall system. Methods for analog ambient backscatter have been also proposed [Smith et al 17] and [Bletsas et al 17]. Such methods provide analog information transmission, while requiring a frequency modulated (FM) ambient carrier to operate according to their intended purpose. 
     SUMMARY OF THE INVENTION 
     The present invention discloses methods and apparatus for ultra-low power wireless transmission and reception of information. Two methods and apparatus for backscattering information when the illuminating radio frequency (RF) signal is a modulated signal, namely pseudo FSK and shifted BPSK, are disclosed. Methods and apparatus for receiving signals resulting from embodiments of the backscattering methods pseudo FSK and shifted BPSK, are also disclosed. 
     Pseudo FSK modulation is designed so as to operate when the (modulated) signal of the ambient illuminating carrier is of constant envelope modulation. In pseudo FSK modulation, wireless channel parameters are assumed constant for the duration of a certain number of bits. Reception of pseudo FSK is fully coherent and estimation of related parameters is performed through short training sequences. 
     Shifted BPSK modulation is designed so as to operate with any (modulated) signal of the ambient illuminating carrier, irrespective of its modulation or structure. An aspect of shifted BPSK also utilizes error correction channel coding. Reception method of channel-coded or channel-uncoded shifted BPSK-modulated backscattered signals, does not need any information regarding the ambient illuminating signal. 
     A short training sequence backscattered from the tag is utilized for acquiring a tag and wireless channel related phase. The methods and apparatus of embodiments of this invention allows for reliable backscatter communication; the existence of modulation at the ambient illuminator is turned to an advantage by this invention. More particularly, reliability of backscatter communication is increased. 
     Both backscatter modulation methods offer a straightforward way for multiple access in the frequency domain. Multiple access in the frequency domain can be exploited for networking purposes, without requiring receivers or codes at each tag. 
     Thus, multiple access from several tags is possible without TDMA or CDMA methods. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a setup for utilization of an embodiment of part of the invention. 
         FIG. 2  shows an embodiment of part of the invention, a backscatter tag. 
         FIG. 3  shows waveforms produced by an embodiment of the backscatter tag. 
         FIG. 4  shows an embodiment of another part of part of the invention, a receiver. 
         FIG. 5  shows a performance of an embodiment of the receiver for pseudo FSK modulation. 
         FIG. 6  shows performance of embodiments of receivers for pseudo FSK and shifted BPSK. 
         FIG. 7  shows a performance of an embodiment of the receiver for uncoded shifted BPSK. 
         FIG. 8  shows a performance of an embodiment of the receiver for coded shifted BPSK. 
         FIG. 9  shows signal jamming due to operation of an embodiment of the backscatter tag. 
         FIG. 10  shows an embodiment of a method for radio frequency (RF) source localization. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     An illuminator  100  transmits a modulated signal, destined to its own, legacy receiver ( FIG. 1 ). The same signal illuminates the tag antenna  205  in  FIG. 1 ; the tag modulates its own information on the already modulated illuminating signal, by careful reflection. The tag antenna-reflected signal is received by the receiver (Rx)  400 , through the receiver antenna  490  ( FIG. 1 ). 
     One embodiment of part of the invention, the backscatter tag, is shown in  FIG. 2 . Backscatter tag consists of an antenna  205  connected to a radio frequency (RF) termination switching module  210 . The RF switching module consists of a RF switch  211  and two complex impedances  212  and  213 . Antenna  205  is connected to either complex impedance (load) Z 0    212  or complex impedance (load) Z 1    213  by RF switch  211 . 
     Load connected to the tag antenna, at any given time instant, is selected based on the value of a binary signal  221 , driving the RF switch. Binary signal  221  is the output of a multiplexer  220 . Based on a signal  242 , the output  221  of the multiplexer  220  is a signal  241  or a signal  261  or a signal  271 . Only one of signals  241 ,  261 ,  271  can be assigned to signal  221  at a specific time instant. 
     In the step of producing pseudo FSK ( FIG. 3 , left waveforms) backscatter modulation, a Processing Unit  240  drives multiplexer  220  via the signal  242 . Processing Unit  240  drives signal  242  such that signal  241  is assigned to signal  221 . In this step, RF switch  211  alternates the loads according to signal  241 , produced by Processing Unit  240 . 
     When bit  1  is to be backscattered under pseudo FSK backscatter modulation, a square wave of frequency F sw  is generated by Processing Unit  240  for the duration of the bit. The square wave produced in Processing Unit  240  is assigned to signal  241 . That way RF switch  211  alternates between the load  212  and the load  213 , for the duration of the bit at an alternation rate of F sw . The 2 levels of the produced square wave correspond to the levels accepted by the RF switch  211 . 
     When bit  0  is to be backscattered (under pseudo FSK), Processing Unit  240  assigns to signal  241  a direct current (DC) waveform. The direct current (DC) waveform attains either the high or low state for time duration equal to the bit duration. Such high or low state is the same as the high or low state of the square wave produced for backscattering bit  1 . That way the RF switch  211  selects one of the two termination loads for the duration of bit  0 . An example of signal  241  is depicted at  FIG. 3  left and top. 
     In the step of producing shifted BPSK ( FIG. 3  right waveforms), Processing Unit  240  assigns to signal  242  a value such that the signal  271  is assigned to the signal  221  via the multiplexer  220 . In this step, RF switch  211  alternates the loads according to signal  271 , produced by a BPSK Modulator  270 , explained in detail below. 
     Under shifted BPSK modulation, the BPSK Modulator  270  produces signal  271  depending on a signal  245 . In the embodiment in  FIG. 2 , signal  245  is produced by Processing Unit  240  based on the bits to be backscattered. Signal  245  is a square pulses train, with the level of each pulse corresponding to the bit to be backscattered, at the specific time instant. Processing Unit  240  is able to choose whether or not to utilize an error correction code on the bit sequence to be assigned on signal  245 . BPSK Modulator  270  produces a square wave (which is assigned to signal  271 ) of frequency F sw , with its phase depending on the value of signal  245 . Phases introduced in the embodiment of the present description are 
               ϕ   1     =     π   2           
when signal  245  dictates backscattering bit  1  and ϕ 0 =0 when bit  0  is to be backscattered. In another embodiment, BPSK Modulator is adjustable, offering other values for produced phases. Phase
 
               ϕ   1     =     π   2           
is mapped to the square wave by introducing an initial time shift equal to a quarter of the inverse F sw  to the said square wave. Phase ϕ 0 =0 is mapped by not introducing an initial phase. An example of signal  271  is depicted in  FIG. 3  right and top.
 
     Embodiment in  FIG. 2  facilitates multiple sensors denoted by Sensors block  280 . Each sensor may be active or passive. Capacitive and resistive sensing elements, whose characteristic value (i.e., capacitance or resistance, respectively) changes with respect to a measured quantity, constitute passive sensors. Passive sensors do not require external supply to translate a measured quantity to an inherent electrical characteristic (resistance, capacitance, voltage, current etc.). Sensing elements that require some form of power supply to translate a measured quantity to a measurable electrical characteristic (resistance, capacitance, voltage, current etc.), constitute active sensors. Sensors block  280  is considered to facilitate passive or active signal conditioning modules for interfacing the passive or active sensors. 
     For passive sensing elements (resistive or capacitive), interfacing with the Processing Unit  240  is done through a resistor/capacitor (RC) oscillator  290 . RC oscillator  290  produces a periodic wave of which the frequency depends on the values of the passive sensing element(s), part of Sensors block  280 . Periodic wave produced by RC oscillator  290  is assigned to a signal  291  and a signal  292 . Processing Unit  240  performs frequency counting on signal  291  and computes the value measured by the passive sensing element(s). Processing Unit  240  arranges the necessary bit sequence, corresponding to the measured value and performs the preferred (pseudo FSK or shifted BPSK) backscattering operation. 
     Values of passive sensors can be also backscattered in an analog manner utilizing principles shown in [Bletsas et al 17]. In the embodiment in  FIG. 2 , such an operation is done through a multiplexer  250  and a voltage controlled oscillator (VCO)  260 . Processing Unit  240  assigns to a signal  243  a value such that the multiplexer  250  assigns to a signal  251  the signal  292  from RC oscillator  290 . Processing Unit  240  then assigns to signal  242  a value corresponding to multiplexer  220  assigning to signal  221  the signal  261 . That way the RF switch  211  alternates between the loads  212  to  213 , based on the output  261  of voltage controlled oscillator (VCO)  260  and the operation of [Bletsas et al 17] is obtained. 
     In another implementation example, Processing Unit  240  assigns to the signal  243  a value such that the multiplexer  250  assigns to signal  251  a signal  244 . That way, Processing Unit  240  can directly assign via the signal  251 , values to the voltage controlled oscillator (VCO)  260 . This operation allows for Processing Unit  240  to encode information to be frequency modulated by the voltage controlled oscillator (VCO)  260 . 
     For active sensors facilitated in Sensors block  280  of the embodiment in  FIG. 2 , an analog to digital converter (ADC)  200  is utilized. After signal conditioning is performed on Sensors block  280 , offering a voltage value corresponding to a measured quantity, the analog to digital converter (ADC)  200  measures the voltage through signal  281 . Processing Unit  240  then receives samples from the analog to digital converter (ADC)  200  through a signal  201 , arranges the appropriate bit sequence and backscatters using the preferred (pseudo FSK or shifted BPSK) backscattering operation. 
     In the embodiment in  FIG. 2 , a Power Management system  230  is considered. Power Management system  230  supplies power to all the components in the system, using signals  231  and  232 . Processing Unit  240  communicates with Power Management system  230  using signal  246 . Processing Unit  240  may control Power Management system  230  so as to selectively power off parts of the embodiment in  FIG. 2  when not used at a specific time instant. 
     Power Management system  230  may harvest power from various ambient sources. In the embodiment described, no specific harvesting source is considered. In another embodiment, specific harvesting elements e.g., solar panels or radio frequency (RF) energy harvesting circuitry can be utilized. 
     Processing Unit  240  must select which backscattering method will be employed to backscatter a specific number of bits. The selection is made between 2 digital methods, pseudo FSK and shifted BPSK. A third method is also available in the embodiment in  FIG. 2 , based on communication principles presented in [Bletsas et al 17]. 
     When the modulation structure of ambient illuminating carrier  100  ( FIG. 1 ) attains the form of a constant envelope modulated signal and wireless channel parameters remain constant for the duration of a backscattered packet, Processing Unit  240  will be adjusted so as to utilize pseudo FSK. In the case when no information about the ambient carrier is available, Processing Unit  240  will utilize shifted BPSK. Error correction coding can be also utilized under shifted BPSK modulation. In another embodiment of part of the invention, when ambient illuminating carrier  100  ( FIG. 1 ) emits an unmodulated carrier, either of the methods can be utilized. 
     Other modulation selection criteria may be applied. Available energy based selection may be utilized when small amounts of energy are available through Power Management system  230 . 
     One embodiment of a second part of the invention, a receiver  400 , is shown in  FIG. 4 . Embodiment of the receiver shown in  FIG. 4  includes components that extract information from the signal backscattered from tag of the embodiment in  FIG. 2 . 
     An antenna  490  is connected through  411  to the receiver  400 . An RF Front-End  410  is utilized. The RF Front-End  410  is utilized in order for band pass filtering around a specific frequency to be implemented and an initial downconversion to an intermediate frequency (IF) analog signal  412  to be achieved. The analog signal  412  is then sampled by an analog to digital converter (ADC)  420  at a rate required for handling the bandwidth of the analog signal  412 . The sampled, digitized version of the analog signal  412  is a signal  421 . For homodyne reception, the signal  421  is further downconverted using a digital downconverter (DDC)  440 . In another embodiment, the RF Front-End  410  utilizes mixers that downconvert the incoming signals from the antenna  490  through signal  411 , directly to baseband (homodyne reception). RF Front-End also includes low pass filters to limit the spectral components of the downconverted signal within a desired baseband bandwidth. In such an alternative embodiment, use of the digital downconverter (DDC) is omitted. 
     After sampling and downconversion to baseband, a digital signal processing (DSP) module  450  implements signal processing methods in order for the backscattered bits to be recovered. After recovering the bits, an Interface module  460  handles the communication with an outside entity. A smartphone or a computer may constitute outside entities. In the embodiment in  FIG. 4 , a computer (PC)  480  may be used to acquire and/or visualize the recovered information. A Custom Interface  470  is part of the receiver to acquire/visualize the recovered data without the need of a computer (PC)  480 . The communication between Interface module  460 , digital downconverter (DDC)  440  and digital signal processing (DSP) module  450 , is controlled by a microprocessor unit (MPU)  430 . Signals  422  and  411  are used by the microprocessor unit (MPU)  430  to control RF Front End  410  and analog to digital converter (ADC)  420  operation. In other embodiments, MPU  430  and DSP  450  modules are not distinct entities. 
     In the case where ambient illuminating carrier  100  ( FIG. 1 ) provides a constant envelope modulated signal and the wireless channel parameters do not change for the duration of a backscattered packet, coherent detection can be applied. Two correlators with complex exponential basis functions are implemented in the digital signal processing (DSP) module  450 . Correlators use samples corresponding to the duration of the bit to be detected. Correlator operation for appropriate bit-level synchronization is equivalent to match filtering. One correlator is used for frequency F sw  and another for −F sw . 
     Denoting as r a vector output of the 2 correlators, corresponding to the received, filtered signal for one bit, a coherent maximum likelihood (ML) detection rule is ∥r−μ r ∥ 2   2   ∥r∥ 2   2 . 
     μ r  is a compound parameter that is assumed unchanged for the duration of the backscattered packet. Parameter μ r  includes tag and wireless channel related parameters. Additionally, μ r  includes statistics of the signal of the ambient illuminating carrier. Rule ∥r−μ r ∥ 2   2   ∥r∥ 2   2  is implemented in the digital signal processing (DSP) module  450 . It should be noted that the expressions assume perfect synchronization and carrier frequency offset (CFO) correction. 
     Parameter μ r  can be estimated using a short training sequence. Training sequence is known to both the Processing Unit  240  of the backscatter tag of the embodiment in  FIG. 2  and the digital signal processing (DSP) module  450  in the embodiment in  FIG. 4 . The estimation is done through a least squares (LS) estimator or a linear minimum mean squared error (LMMSE) estimator, implemented in digital signal processing (DSP) module  450 . 
     The performance of an embodiment of the receiver for pseudo FSK with respect to bit error rate (BER), is shown in  FIG. 5 . It can be observed that an analytical bit error rate (BER), closed form expression corresponding to the detection rule ∥r−μ r ∥ 2   2   ∥r∥ 2   2 , matches simulation results, under perfect channel state information (CSI). It can also be seen that under perfect channel state information (CSI), compared to N tr =4 (N data =96), the maximum likelihood (ML) detection rule offers ˜2 dB better performance than using the estimated channel. Allocating more bits for channel estimation purposes, lowers the difference between the maximum likelihood (ML) detection rule with perfect channel state information (CSI) and the maximum likelihood (ML) detection rule using the channel estimate. Specifically, for the chosen values and for 6 more training bits (6 less data bits/packet) the difference reduces from ˜2 dB to ˜1 dB. 
     The simulation results are provided under unit power Rayleigh fading for all involved wireless channels. The packet has a fixed length of N tr +N data =100 bits. The signal of the ambient illuminating carrier was modeled as a constant envelope signal with the information modeled as a zero mean Gaussian process. Channel state information (CSI) includes all the unknown parameters, including the statistics of the signal of the ambient illuminating carrier. In pseudo FSK case, full and perfect channel state information (CSI) is the state of having full and perfect knowledge of parameter μ r . 
     In the case of shifted BPSK, the receiver requires no information regarding the signal of the ambient carrier nor its structure. The same correlators as in pseudo FSK are used in the digital signal processing (DSP) module  450 . The detection rule under shifted BPSK modulation is cos(2Φ t +θ p,n ) 0. Phase Φ t  is a random phase (which is considered constant for the duration of the backscattered packet) introduced by tag&#39;s operation and wireless channels. (r s,n   + ), (r s,n   − ) are the outputs of the two correlators for the n-th bit. The following complex number |r p,n |e jθ     p,n   =(r s,n   + )*(r s,n   − ) can be defined for the n-th tag-backscattered bit, also defining phase θ p,n  for the n-th bit. Operator (.)* stands for conjugating the complex argument. Rule cos(2Φ t +θ p,n ) 0 is applicable when no error correction coding is utilized. It should be noted that the expressions assume perfect synchronization and carrier frequency offset (CFO) correction. 
     In another example, where error correction coding is utilized by the embodiment in  FIG. 2 , a different detection rule is implemented in the digital signal processing (DSP) module  450  of the embodiment in  FIG. 4 . A detection rule under (error correction) coded shifted BPSK is 
                 c   ^     =     arg   ⁢       max     c   ∈   C       ⁢       ∑     n   =   1       N   c       ⁢       w   n     ⁢     c   n               ,         
where C stands for the set of all possible codewords of the utilized error correcting code. Weights w n  are defined as w n =−|r p,n |cos(2Φ t +θ p,n ). It should be noted that the expressions assume perfect synchronization and carrier frequency offset (CFO) correction.
 
     To acquire Φ t , estimation is performed in the digital signal processing (DSP) module  450  using 
                   Φ   ^     t     =       ∠   ⁢           ⁢         r     1   :     N   tr       +     ⁡     (     r     1   :     N   tr       -     )       H       2       ,         
where operator (.) H  stands for taking the conjugate transpose of the complex vector argument. The method utilizes a short sequence of N tr  bits known to both the Processing Unit  240  in  FIG. 2  and the digital signal processing (DSP) module  450  of the embodiment of the receiver in  FIG. 4 . r 1:N     tr     + , r 1:N     tr     −  are row vectors containing the output of the correlators for the N tr  bits of the training sequence. In a different embodiment of part of the invention, other methods (e.g., exploiting some form of differential encoding or other geometrical methods) are possible.
 
     The performance of an embodiment of the receiver for uncoded shifted BPSK, with respect to bit error rate (BER), is shown in  FIG. 7 . Comparison is performed between detection rule cos(2Φ t +θ p,n ) 0 and a maximum likelihood (ML) detection rule utilizing full (and perfect) information regarding channel state information (CSI). Detection rule cos(2Φ t +θ p,n ) 0 is also evaluated for the following cases: 1) available knowledge of Φ t  and 2) estimate {circumflex over (Φ)} t , for N tr =1,5,10. It can be observed that the detection rule cos(2Φ t +θ p,n ) 0 with perfect information regarding Φ t , offers 4 dB worst performance than the maximum likelihood (ML) detection rule utilizing full and perfect channel state information (CSI). 
     When no information about Φ t  is available, it can be seen in  FIG. 7 , that for 10 training bits, the difference between the detection rule cos(2Φ t +θ p,n ) 0 utilizing perfect Φ t  and the same rule using {circumflex over (Φ)} t  instead, is approximately 0.5 dB. When 1 training bit is used, the difference increases to ≈4 dB, resulting a loss of 8 dB compared to the maximum likelihood (ML) detection rule utilizing full and perfect channel state information (CSI). The same channel and packet parameters as with the previous paragraphs were utilized throughout the simulations. Channel state information (CSI) includes all the unknown parameters, including the signal of the ambient illuminating carrier. 
     The performance of an embodiment of the receiver for coded shifted BPSK, with respect to bit error rate (BER), is shown in  FIG. 8 . All depicted cases assume the signal of ambient illuminating carrier to be modulated. For the special case of CIVC explained below, ambient illuminating carrier is assumed constant for the duration of the tag packet, while in all other cases, it changes across consecutive tag bits. It is clear that the (coded) detection rule 
                 c   ^     =     arg   ⁢       max     c   ∈   C       ⁢       ∑     n   =   1       N   c       ⁢       w   n     ⁢     c   n               ,         
outperforms the detection rule when no coding is utilized, cos(2Φ t +θ p,n ) 0. It is also observed that in the high signal-to-noise ratio (SNR) regime, the detection rule
 
               c   ^     =     arg   ⁢       max     c   ∈   C       ⁢       ∑     n   =   1       N   c       ⁢       w   n     ⁢     c   n                   
offers slightly better performance, compared to the maximum likelihood (ML) detection rule utilizing full and perfect channel state information (CSI), when no coding is used.
 
     Two cases are additionally demonstrated in  FIG. 8 . Case where both the signal of the ambient illuminating carrier and the wireless channel parameters vary between successive bits, which will be referred to as varying illuminator varying channel (VIVC) and the case where the signal of the ambient illuminating carrier remains constant during the packet but the channels vary among consecutive bits (constant illuminator, varying channel-CIVC). In the last case, the signal of the ambient illuminating carrier was held constant for the duration of a packet. 
     The performance gain offered when both the signal of the illuminating carrier and the wireless channel parameters vary between successive bits, is the result of the error correcting code being fully utilized. Constant wireless channel parameters during the transmission of multiple bits may introduce correlation between the received statistics. Thus the code may not be able to offer its best performance. In a similar manner, when the signal of the ambient illuminating carrier remains constant for the duration of the packet while the channels vary (CIVC), the same reasoning can be applied.  FIG. 8  shows that modulation at the ambient signal assists the coded detection rule 
               c   ^     =     arg   ⁢       max     c   ∈   C       ⁢       ∑     n   =   1       N   c       ⁢       w   n     ⁢     c   n                   
and radically improves performance, even though the detection rule
 
               c   ^     =     arg   ⁢       max     c   ∈   C       ⁢       ∑     n   =   1       N   c       ⁢       w   n     ⁢     c   n                   
requires minimal information.
 
     Modeling of the signal of the ambient illuminating carrier for the purposes of simulations resulting to  FIGS. 7 to 8 , was done using a proper zero mean complex Gaussian process. For the coded shifted BPSK, BCH (31,11) channel code was utilized. 
       FIG. 6  offers a comparison between the bit error rate (BER) performance of (uncoded) shifted BPSK and pseudo FSK. The pseudo-FSK modulation outperforms shifted BPSK, at the low signal-to-noise ratio (SNR) regime, for approximately 1 dB. As the signal-to-noise ratio (SNR) increases, the gap between the two schemes decreases while for signal-to-noise ratio (SNR) ˜20 dB, the gap vanishes. This behavior may be explained by the fact that the estimation method for Φ t  in the shifted BPSK is a heuristic based on the assumption of the absence of noise. Additionally, the heuristic estimation used in shifted BPSK only estimates Φ t , while the LS estimation used in pseudo-FSK, estimates the compound parameter μ r . Compound parameter μ r , except Φ t , includes tag, wireless channel and ambient illuminating carrier related parameters which are assumed constant for N pack  bits. 
     In the simulations resulting  FIG. 6 , the signal of the ambient illuminating carrier was modeled as a constant envelope signal with the information modeled as a zero mean Gaussian process. 
     Methods pseudo FSK and shifted BPSK also offer advantages when signals from multiple distinct devices, which constitute embodiments of part of the invention in  FIG. 2 , need to be received simultaneously from embodiment of the receiver in  FIG. 4 . 
     Specifically, different F sw  values can be utilized at each distinct device constituting embodiment of part of the invention in  FIG. 2 . That way multiple access in the frequency domain can be achieved and networking is attainable, without using code division (CDMA) or time domain division multiple access (TDMA). 
     An advantageous variation of the embodiment of part of the invention in  FIG. 2  is the one of performing a passive jamming on full duplex transceivers. Processing Unit  240  continuously generates a square wave of frequency F sep . The square wave produced is assigned to signal  241  and the Processing Unit  240  assigns to signal  242  a value such that the multiplexer  220  assigns to signal  221  the signal  241 . 
     The operation described in the previous paragraph, results to the RF switch  211  be driven by a square wave of frequency F sep . 
     A passband signal s(t) centered at F s  whose frequency components are within the bandwidth of both antenna  205  and switching module  210 , is assumed to impinge on antenna  205 . Operation of RF switch  211  will result to the creation of attenuated versions of signal s(t). Specifically, such tag operation results to attenuated versions/copies of signal s(t), appearing at frequencies F s ±F sep . 
     A full duplex transceiver has a frequency separation between an uplink and a downlink frequency band equal to F sep . In an embodiment of an arrangement in  FIG. 9 , downlink band is located F sep  Hz above the uplink band. When the embodiment of part of the invention in  FIG. 2  is located close to the full duplex transceiver and the operation described in the previous paragraphs is utilized, attenuated versions of the transmitted (uplink) signal will appear at frequencies F s ±F sep . 
     Operation of embodiment of part of the invention in  FIG. 2 , results to spectral components of the uplink (of the full duplex transceiver) signal to appear in the downlink band of the full duplex transceiver. If the signal copy received in the downlink band of the full duplex transceiver is not substantially attenuated, jamming will occur at the full duplex transceiver. The full duplex transceiver will not be able to receive information for as long as it transmits. The operation is depicted in  FIG. 9 . 
     Another advantageous variation of the embodiment of part of the invention in  FIG. 2  is the one for aiding RF source localization, as shown in an arrangement in  FIG. 10 . 
     Tag  1  and Tag  2  of the arrangement depicted in  FIG. 10 , are implementations of embodiment of part of the invention in  FIG. 2 . 
     Reader  400  in  FIG. 10 , is an implementation of an embodiment of the receiver  400  in  FIG. 4 . RF source  100  depicted in  FIG. 10 , is a source of a narrowband, modulated RF signal centered at F c  whose location must be estimated. 
     Tags  1  and  2  in  FIG. 10  operate as described in the previous paragraphs with, however different switching frequencies, F sw   1 , F sw   2 . 
     Operation of Tag  1  and Tag  2 , results to the creation of additional wireless paths between the RF source and the reader. 
     Due to operation of Tag  1  and Tag  2 , the signal emitted from RF source is now observable from 5 new frequencies, F c ±F sw   1 , F c ±F sw   2  and F c . 
     Observing signal of RF source  100  from 5 frequencies in total, offers signal diversity in the frequency domain. Such a diversity gain can be exploited by localization methods to achieve an overall increase in performance, with respect to localization accuracy. 
     Embodiment of the arrangement in  FIG. 10  does not need to utilize multi-element antenna array in the Reader  400 . Utilizing an antenna array would increase both the cost and the complexity involved in implementing a receiver of signals. Spatial diversity, potentially offered through utilization of an antenna array, is exchanged with frequency diversity offered by low-cost, ultra-low power embodiments of part of the invention in  FIG. 2 . Placing embodiments of part of the invention in  FIG. 2  at distant from Reader  400  places, offers an increase in the effective antenna aperture via frequency domain diversity. 
     Thus, localization methods can benefit from such an embodiment of the arrangement in  FIG. 10 . 
     While the foregoing written description of the invention enables one of ordinary skill to make and use what is considered presently to be the best mode thereof, those of ordinary skill will understand and appreciate the existence of variations, combinations, and equivalents of the specific embodiment, method, and examples herein. The invention should therefore not be limited by the above described embodiment, method, and examples, but by all embodiments and methods within the scope and spirit of the invention as claimed. 
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