Patent Publication Number: US-7221107-B2

Title: Low frequency electronic ballast for gas discharge lamps

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   Not Applicable 
   BACKGROUND OF THE INVENTION 
   The present invention relates to a low frequency power converter and specifically to low frequency electronic ballasts for gas discharge devices. More specifically, the present invention relates to a low frequency square wave electronic ballast for high intensity discharge (HID) lamps. 
   The prior art is replete with many known circuits providing electronic ballast for gas discharge lamps. For instance, high efficient electronic ballast which can be used with HPS (HID) lamps are discussed in U.S. Pat. No. 5,313,143 entitled “Master-slave half-bridge DC-to-AC switchmode power converter”, and U.S. Pat. No. 6,329,761 entitled “Frequency controlled half-bridge inverter for variable loads”, from the same inventor of the present invention. Furthermore, a low frequency square wave electronic ballast, especially for metal halide (MH) lamps are discussed in U.S. Pat. No. 5,428,268, entitled “Low frequency square wave electronic ballast for gas discharge devices”, also from the same inventor of the present invention. The present invention has several basic differences if compared to the previously mentioned low frequency square wave ballast. 
   Introduction of a new solution for zero current sensing (which is an important functional part for both the input and current source units), a simple temperature compensated nonlinear function generator, the implementation logic supplies for the floating switches of the low frequency full-bridge inverter are among the main improvements and a more effective ignition solution. Further low frequency electronic ballast are discussed in U. S. Pat. No. 5,710,488 entitled “Low-frequency high-efficacy electronic ballast”, from Nilssen, U.S. Pat. No. 4,614,898 entitled “electronic ballast with low frequency AC to AC converter” from Itani et al, 1986, U.S. Pat. No. 6,166,495 entitled “square wave ballast for mercury free arc lamp”, from Newell et al, and U.S. Pat. No. 5,235,255 entitled “Switching circuit for operating a discharge lamp with constant power” from Blom. Still further advantages of the present invention comparing to mentioned patent applications will become apparent from a consideration of the ensuing description and drawings. 
   An important application for high frequency switchmode power converters is supplying power to gas discharge devices, especially high intensity discharge (HID) lamps. Therefore, the efficiency of the conventional core&amp;coil ballast can be significantly improved and the weight decreased. In the case of high frequency powering of gas discharge lamps, the high frequency ballast and the gas discharge lamp have a higher level of interaction than that which exists between a conventional low frequency ballast and gas discharge lamp. High frequency ballasts, where the frequency of lamp current higher than 4 kHz, may suffer from acoustic resonance which can cause various problems such as instability, high output fluctuation, or, in the worst case, cracked arc tubes. Therefore, an optimum solution to this problem is the use of a high frequency DC-to-DC switch-mode converter as a controlled current source connected to a low frequency DC-to-AC square wave inverter supplying the gas discharge lamp. Due to its lessened weight, higher efficiency and the nonexistence of flickering and acoustic resonances, this novel high frequency ballast providing low frequency square wave current for the HID lamps, has significant advantages when compared with either the conventional low frequency ballasts and the usual high frequency electronic ballast. Additionally, a new, high sophisticated electronic ballast generation can be introduced to provide several special features, such as, for example, automatic or controlled dimming providing significant energy saving in a wide temperature range. 
   BRIEF SUMMARY OF THE INVENTION 
   It is an object of the present invention to provide an acoustic resonance and flickering free, high efficient low frequency square wave electronic ballast for high intensity gas discharge lamps operating in wide temperature range providing extended operational life time and energy saving. 
   A second object of the present invention to provide a dimmable electronic ballast for high intensity gas discharge lamps providing further energy saving. 
   A further object of the present invention to provide a high power factor input unit implementing a DC power supply for electronic ballast, wherein no electrolytic capacitors are used; 
   Another object of the present invention to provide a DC current source, wherein the output power can be externally controlled in a given range implementing dimming, wherein no electrolytic capacitors are used; 
   Further object of the present invention to provide a floating logic control circuit controlling a high frequency buck converter as a DC current source; 
   Another object of the present invention to provide a highly efficient square wave full-bridge inverter operating in a very wide frequency range including DC operation, wherein no electrolytic capacitors are used; 
   Further object of the present invention to provide a logic control circuit controlling a square wave full-bridge inverter implementing transition between the high (or zero) and the low frequency operations; 
   Another object of the present invention to provide a high frequency, high voltage ignition solution for reliable ignition of HID lamps. 
   Further object of the present invention to provide ideal ballast curve for HID lamps, wherein the lamp power is independent from the line voltage fluctuation and the lamp voltage increasing during the lamp life time; 
   These and other objects, features and advantages of the present invention will be more readily apparent from the following detailed description, wherein reference is made to the drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings, closely related figures have the same numbers but different alphabetic suffixes. 
       FIG. 1A  illustrates the block diagram of preferred electronic ballast for gas discharge lamps; 
       FIG. 1B  shows the output voltage wave form of the Input Unit and the rectified input voltage 
       FIG. 1C  illustrates the output voltage and current of the Current Source. It also shows the minimum level of its input voltage. 
       FIG. 1D  shows the square wave lamp voltage and lamp current. 
       FIG. 1E  illustrates the diagram of lamp current vs. lamp voltage and the preferred ballast curve 
       FIG. 2A  shows the circuit diagram of the Input Unit and its Control Unit. It also shows the Interface Unit and the Logic Supply. 
       FIG. 2B  illustrates the current wave form of the main switch T 1  and its control signal. 
       FIG. 2C  shows the current and voltage wave forms of rectifier D 2  shown in  FIG. 2A . 
       FIG. 2D  shows the detailed circuit diagram of the Interface Unit, providing external dimming and ON/OFF control. 
       FIG. 2E  illustrates the detailed circuit diagram of the Control Unit of the preferred Input Unit. 
       FIG. 3A  illustrates the circuit diagram of the DC Current Source; 
       FIG. 3B  shows the detailed circuit diagram of the Control Unit of the preferred DC Current Source shown in  FIG. 3A ; 
       FIG. 3C  shows the basic wave forms of the preferred DC Current Source and its Control Unit. 
       FIG. 4A  shows the circuit diagram of a Square Wave Inverter designated as the Output Unit in  FIG. 1A  and its Control Unit. 
       FIG. 4B  shows the detailed circuit diagram of the Timer/Comparator subunit of the preferred Control Unit of the Square Wave Inverter; 
       FIG. 4C  shows the detailed circuit diagram of the Logic Driver/Oscillator subunit of the preferred Control Unit of the Square Wave Inverter; 
       FIG. 4D  shows the basic wave forms of the Current Limiter subunit of the preferred Control Unit of the Square Wave Inverter; 
       FIG. 4E  shows the detailed circuit diagram of the Current Limiter subunit of the preferred Control Unit of the Square Wave Inverter; 
       FIG. 4F  illustrates the HF to LF transition from circuit topological view. 
       FIG. 4G  shows the basic current and voltage wave forms with respect to the high frequency (HF) to low frequency (LF) transition. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Generally, the high frequency electronic ballasts have shown limitation factors which severely restrict the availability of commercial applications for the HID lighting industry. Due to the fact that acoustic resonance is produced in a variety of different frequency ranges, which ranges are themselves dependent upon the lamp characteristics. In other words, a high frequency electronic ballast will cause acoustic resonance in some HID lamps, but not in others. Naturally, this draw-back makes it impossible to market a universally acceptable electronic HID ballast which may be used with any lamp other than a lamp with which the ballast has been specifically tested, in order to ensure that their is no acoustic resonance. 
   For overcoming the disadvantages of the high frequency electronic ballasts, an electronic ballast having high efficiency (≈95%) and low frequency square wave output current is suggested as illustrated in  FIG. 1A  including the main three units of the preferred low frequency square wave electronic ballast, namely:
         an Input Unit, including a power factor preregulator, an interface circuit for external control, and logic supply providing stabilized 12V for the all control units of the ballast. The output voltage of the Input Unit (V 1 ) and the rectified input voltage (Vi) are shown in  FIG. 1B  where the power factor Preregulator is based on a boost converter configuration;   a Current Source, which can be considered as a voltage to current converter implementing the ideal ballast curve shown in  FIG. 1E . In this case, the current in low output voltage (0&lt;20V) can be lowered, but it should be sufficiently high, forcing the transition from glow discharge to arc discharge at a certain glow discharge voltage determined by the lamp.  FIG. 1C  shows the output voltage and current levels determined by the lamp, if the current source is based on a buck converter configuration (V 1 &gt;V 0 );   an Output Unit (full-bridge inverter), as a solution to the acoustic resonance problem caused by high frequency lamp current, low frequency (50 Hz-500 Hz) square wave lamp current is implemented as it is shown in  FIG. 1D . In the case of a low frequency square wave lamp current, the temperature modulation of the central discharge channel is almost zero. However, since the polarity change of the lamp current is not instantaneous, especially if a low inductance ignitor transformer is connected in series with the lamp, the lamp power fluctuates twice of the current frequency. Since the transition is very fast (&lt;10 μs) with respect to a half-period (5-10 ms), the flickering is negligible. Also, for the same reason, the high frequency harmonics of the lamp current are significantly smaller than in the high frequency case.       

   From electronic circuit viewpoint a square wave ballast is more complex than a simple high frequency inverter. It should contain at least two power unit, namely a power controlled current source and a low frequency full-bridge inverter. Furthermore, if high power factor is required, it should be also included a high power factor pre-regulator. Therefore, the increased complexity and higher cost of a low frequency square wave electronic ballast may restrict its industrial application to areas where special requirements are demanded, namely extremely wide temperature range and flickering free operation. Special circuit solutions for overcoming the technical barriers from ballast and electronic circuit viewpoints will be presented in the following detailed descriptions. 
   Input Unit 
   The overall efficiency and the cost of an electronic ballast device is crucial. Therefore, only a simple but very highly efficient (&gt;97%) circuit solutions can be considered still providing high power factor and low total harmonic distortion. Since a simple rectifier and filter can produce large third harmonic distortion and the power factor is extremely low (&lt;50%), application of a high power factor input unit (pre-regulator) is required. In this case the relative simplicity and very high efficiency can be considered as the main design goals. From industrial application viewpoint the very low THD (&lt;3%) and the ideal power factor(100%) are not required. An acceptable compromise is: THD&lt;10% and PF&gt;97%. According to these requirement, as it is shown in  FIG. 2A , a boost converter configuration in discontinuous border mode can be considered as the optimum solution even if the amplitude of the inductor current is higher then in continuous mode. In this case, the zero current switching, especially at higher voltages (200V-400V) dramatically decreases the stress of the switches, therefore increasing the reliability and efficiency of the overall circuit. In  FIG. 2A  the main components of boost converter—connected to the Input Filter—are the Inductor L 2 - 1 , MOSFET T 2 - 1 FIG, Rectifier D 2 - 1 , and Capacitor V A . The DC voltage V 21  is proportional to the average value of input voltage. Rectifier D 2 - 2  provides zero current sensing when T 2 - 1  is OFF.  FIG. 2A  also shows the Interface Unit providing isolated dimming and ON/OFF external control. Furthermore, a Logic Supply Unit providing stabilized 12V for the control units of the ballast is also illustrated in  FIG. 2A . 
     FIG. 2B  illustrates the current wave form of the main switch implemented by power MOSFET T 2 - 1  and its gate control signal V 22 .  FIG. 2C  shows the inductor current I 21  in the discontinuous border mode, and the voltage signal V 25  on rectifier D 2 - 2  providing a simple and effective (low power loss) solution for the zero current sensing of the inductor L, where no shunt resistor is applied. Therefore, using a simple comparator (see IC 2 - 11  in  FIG. 2E ), the zero/nonzero values of the inductor current can be easily converted to digital signal. Controlled On-time and zero current switching on techniques are applied. Therefore, the peak and average inductor current is sinusoidal as is the input voltage. Furthermore the control of the circuit in discontinuous mode, based on the constant On Time method, can be easily implemented (no right plane zero) increasing the reliability and efficiency of the overall circuit. 
     FIG. 2D  shows the circuit diagram of the Interface Unit based on comparators IC 2 - 1  and IC 2 - 2 . The whole Interface Unit is isolated from the main part of the ballast (therefore, from the line) and the control connection is implemented by optoisolators OC 2 - 1  and OC 2 - 2 . The dimming(E 1 -E 3 ) can be externally controlled by a simple low power switch (DIM) as it is shown in  FIG. 2A . The ON/OFF control(E 1 -E 2 ) can be also realized by a low power switch, or if it is required, with a photoconductive cell (PR). 
     FIG. 2E  shows the detailed circuit diagram of the preferred Control Unit of the Input Unit including:
         (a) an error amplifier IC 2 - 8  controlling the output voltage V A ;   (b) a sawtooth generator implemented by a resistor R 2 - 1  (R 2 - 1 ×R 2 - 2  in case of dimming controlled by low power MOSFET T 2 - 2 ), a capacitor C 2 - 2 , a low power MOSFET T 2 - 3  and a NAND Schmitt-trigger IC 2 - 10 ;   (c) an ON-time controller implemented by comparator IC 2 - 9 , where the inputs are connected to the sawtooth generator and the error amplifier IC 2 - 8  where the maximum on-time is limited by Zener diode Z 2 - 1 ;   (d) a zero current sensing comparator IC 2 - 11  connected to the rectifier D 2 - 2  and an approximately 4000 mV voltage source;   (e) the voltage comparators IC 2 - 3  and IC 2 - 4  are controlled by voltage V 21  which is proportional to the average value of the rectified input voltage V i , and voltage comparators IC 2 - 5  and IC 2 - 6  are controlled by the output voltage (V A ) of the boost converter;   (f) a temperature controller is implemented by voltage comparator IC 2 - 7  controlled by thermistor TH 2 - 1 ;   (g) a dual input NOR gate controlling the MOSFET Driver of T 2 - 1  ( FIG. 2A ), where the inputs are connected to the zero current sensing comparator IC 2 - 11  and the ON-time controller comparator IC 2 - 9 .       
   An essential difference between the preferred high power factor preregulator of the present invention and standard regulators, is the zero current sensing. In this case, the voltage drop on rectifier D 2 - 2  is compared to the zero level of the control unit providing sensitivity and less loss. This solution is effective if the main switch (T 2 - 1 ) is switched on at zero inductor current level as in the preferred embodiment. A further difference between the preferred high power factor preregulator and standard regulators, is the utilization in the present invention, of a relatively small value film capacitor (C 2 - 1 ) instead of employing a large value electrolytic capacitor as the output capacitor. In the case, the fluctuation (120 Hz) of the output voltage V A  is large as it is illustrated in  FIG. 2B . 
   Current Source 
   With the exception of boost derived converters, several converter configuration may applied as the current source. It can be seen that a basic buck converter as the current source of the low frequency square wave ballast may be an obvious choice, shown in  FIG. 3A . Avoiding extra stress and loss in the switches (T 3 - 1 , D 3 - 1 ), discontinuous border mode for the inductor current I 31  is chosen as it is shown in  FIG. 3C . In this case, the known stability problems of the continuous mode are avoided and a special control method can be applied as the preferred solution.  FIG. 1E  shows the required output power and current vs. output voltage characteristics as the ideal ballast curve for HPS (HID) lamps. The minimum and maximum output voltages are determined by the nominal lamp voltages (100V/55V for HPS, and 130V for MH lamps). 
   The applied control method is significantly different from the usual ones as it will demonstrated in the following part. The control unit, shown in  FIG. 3A , is connected directly to the MOSFET—Driver and therefore to the main switch T 3 - 1 . 
   The zero current sensing of the inductor current I 31  implemented by a fast rectifier D 3 - 2  connected in series with a Schottky-rectifier D 3 - 3  which rectifiers are connected in parallel with the main rectifier D 3 - 1 . If the main switch T 3 - 1  is OFF, the main rectifier D 3 - 1  is ON and an approximately 200 mV voltage drop occurs across the Schottky-rectifier D 3 - 3 . This voltage controls a voltage comparator IC 3 - 3  ( FIG. 3B ) connected to an input of NAND Schmitt-trigger IC 3 - 2 , which forces T 3 - 1  OFF, and allowing the ON state of the main switch T 3 - 1  at zero inductor current. 
   The mapping of inductor current  131  in the ON state of the main switch T 3 - 1  is implemented by rectifier D 3 - 4  connected in series with resistor R 3 - 1  providing charge current for capacitor C 3 - 3 . Therefore, the voltage (12-V 37 ) is proportional to the inductor current I 31 , since both the inductor current and the capacitor voltage V 37  depend linearly on the same voltage: V A -V 0 . Therefore, the peak inductor current as well as the average inductor current can be directly controlled by a reference voltage V 38  ( FIG. 3B ). The discharge of the capacitor C 3 - 3  is achieved by a low power p channel MOSFET T 3 - 2  controlled by the zero current sensing voltage comparator IC 3 - 3  shown in  FIG. 3B . 
   The control of output power can be achieved by implementing the proportionality of the reference voltage V 38  to the inverse value of output voltage V 0 . Therefore, the control of the constant output power can be solved in a certain range of output voltage. Generally, for HID lamps, this output voltage range is: 80V-160V. Continuous dimming of the output power (lamp power) can be achieved by a continuous decrease of the value of resistor R 3 - 1 . The output power can be changed in discrete steps by the values of capacitor C 3 - 3 .  FIG. 3B  shows a solution for this case, where a second capacitor C 3 - 4  is connected parallel with C 3 - 3  controlled by a low power MOSFET T 3 - 3  via an optocoupler OC 3 - 2  providing isolation. Actually, in this case, the full power is provided when MOSFET T 3 - 3  is ON, and dimmed operation if MOSFET T 3 - 3  is OFF. Dimming can be advantageous from an energy saving consideration if the decreased light level is acceptable in certain situations. 
   The electronic realization of the required inverse relationship is implemented by a nonlinear Function Generator shown in  FIG. 3B , based on resistors R 3 - 2 , R 3 - 3 , R 3 - 4 , R 3 - 5 , and diode D 3 - 4 . The output voltage V 0  boosted to the floating control level by rectifier D 3 - 6  and a smoothing capacitor C 3 - 1  as it shown in  FIG. 3A  providing the appropriate voltage level for the function generator. 
   The voltage comparator IC 3 - 4  controls the ON time of the main switch T 3 - 1 . The dual input NOR gate IC 3 - 1  is controlled by the voltages V 33  (V 32  and V 34 ) and V 35  (V 36 ), and its output is connected to the MOSFET Driver shown in  FIG. 3A . 
   The output voltage V 0  is limited by applying a Zener diode Z 3 - 1  connected in series with the optocoupler OC 3 - 2  providing OFF-state for the main switch T 3 - 1 . The corresponding signal wave forms of the circuit diagrams of figures  FIG. 3  A and  FIG. 3  B are illustrated in  FIG. 3C . 
   Output Unit 
   HID lamps are usually supplied (avoiding cataphoretic phenomenon) with symmetrical AC current. Therefore, a symmetrical (D=50%) square wave inverter should be connected to the DC current source including high voltage ignitor circuit. Since the nominal frequency of the inverter is low (50 Hz-500 Hz), only the full-bridge configuration can be considered as it is shown in  FIG. 4A  including a Square Wave Inverter and its Control Unit. The inverter should also operate at high frequency for limited time (≈4s) periodically when the lamp start-up requires increased voltage. 
   Therefore, the application of MOSFET&#39;s are recommended as the main switches (S 1 , S 2 , S 3  and S 4 ), requiring appropriate drivers (DR 1 , DR 2 , DR 3  and DR 4 ). The supply voltages are boosted by rectifiers D 4 - 1  and D 4 - 2  to capacitors C 4 - 3  and C 4 - 4  respectively, wherein their cathodes are connected to capacitor C 4 - 5  charged by 12V Logic Supply. For instance, C 4 - 3  is charged when S 1  is switched on. For ignition purposes, a small pulse transformer TR 4 - 1  is connected in series with lamp. At low frequency, the effect of the transformer can be neglected except for a short time at switching points. The high frequency harmonic components of the lamp current is much lower than at high frequency operation. It follows that the instantaneous power is constant, similarly to the DC operation, except at the switching points, where it goes to zero in a short time interval (≈15 μs). The inductance of the secondary side of the ignition transformer TR 4 - 1  can be utilized for short circuit protection. In this case the peak current can be controlled by a simple circuit, as the current is converted to a proportional voltage signal by resistor Rs.
         (A) T IMER AND  C OMPARATOR . The maximum output voltage range is determined by the current source(0&lt;V 0 &lt;200V). Inside this range the load (lamp) determines the output voltage. When the voltage of an aging lamp achieves approximately 160V, the lamp should be switched off after a certain time delay (12 min.). Furthermore, there should be another (≈170V) voltage level, where the output unit start to operate at high frequency providing sufficiently high ignition voltage for the lamp. Sensing of these two voltage level and converting into digital signals, based on a dual comparator IC 4 - 1  (controlled by V 0 ); is implemented by the Comparator unit shown in  FIG. 4B . If V 0 &lt;160V, V 41 =V 42 =12V. When V 0 &gt;160V, the signal V 41 =0, and when V 0 &gt;170V, the signal V 42 =0. The Timer unit, controlled by signal V 41 , is also shown in  FIG. 4B , including a ripple counter (IC 4 - 2 ) connected to a simple oscillator based on the Schmitt-trigger IC 4 - 3 , a dual input AND-gate IC 4 - 4 , and a monostable multivibrator controlled by signal V 46 . The inverted output  14  of the ripple counter IC 4 - 2  and the output of the monostable multivibrator are AND-gated resulting signal V 44 . After a predetermined time (approximately. 12 min.), the output signal V 44  becomes zero, therefore the inverter will be stopped (see  FIG. 4C ). Selected outputs of the ripple counter (in our case  5 ,  6 , and  7 ) are OR-gated to resistor R 4 - 2  providing the output signal V 43 . As we shall see, the frequency (high or low) of the full-bridge inverter (therefore, the lamp current) is controlled by V 43 .       

   With respect to the output voltage V 0 , the operation of the Output Unit can be summarized as follows:
         1. V 0 &lt;160V→Low frequency operation;   2. 160V&lt;V 0 &lt;170V→Low frequency operation, Timer starts;   3. V 0 &gt;170V→High frequency operation.       

   As we shall see later, when the output voltage decreases to a certain low value (&lt;10V), indicating short circuit, within a short time the Output Unit and the Current Source will be switched off (see Current Limiter) implementing special short circuit protection for the ballast.
         (B) D UAL  F REQUENCY  O SCILLATOR AND  D RIVER . The Dual Frequency Oscillator, shown in  FIG. 4C , provides symmetrical square wave voltage signal V 45  (see output Q). The high frequency (HF) or low frequency (LF) operation of the Dual Frequency Oscillator is controlled by signal Y, where       

   
     
       
         
           Y 
           = 
           
             
               
                 
                   V 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   42 
                 
                 + 
                 
                   V 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   43 
                 
               
               _ 
             
             = 
             
               { 
               
                 
                   
                     
                       1 
                       → 
                       
                         HF 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         operation 
                       
                     
                   
                 
                 
                   
                     
                       0 
                       → 
                       
                         LF 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         operation 
                       
                     
                   
                 
               
             
           
         
       
     
   
   In practice, the low frequency range can be 50 Hz-200 Hz. Lower then 50 Hz can cause flickering as the cataphoretic phenomenon starts to occur. The high frequency range can start at 20 KHz. Essentially higher frequency is not recommended because the increased switching losses. Since the inverter also operates at high frequency as the lamp needs increased voltage at start up, relatively powerful MOSFET drivers should be applied. The MOSFET derivers (DR 1 , DR 2 , DR 3  and DR 4 ) are controlled by driver signals Q 1 , Q 2 , Q 3  and Q 4 , provided by the Driver subunit is also shown in  FIG. 4C . The Driver includes a quad, dual input AND gate IC 4 - 6 . The upper MOSFET drivers DR 3  and DR 4  should include optoisolators having relatively long delay times (&gt;1 μs). Therefore, avoiding the cross conductions of the main switches (S 1 -S 4 , S 2 -S 3 ), the driver signals Q 3  and Q 4  should be delayed according to Q 2  and Q 1 . The delay time (2 μs-5 μs) for the upper switch S 3  (signal Q 3 ) can be adjusted by R 4 - 3  and C 4 - 6  as it is shown in  FIG. 4C . Similarly, the delay time (2 μs-5 μs)for upper switch S 4  (signal Q 4 ) can be adjusted by R 4 - 4  and C 4 - 7  as it is also shown in  FIG. 4C 
         (C) C URRENT  L IMITER . The Current Limiter unit, shown in  FIG. 4E , includes the low voltage comparators IC 4 - 12  and IC 4 - 13 , where the inverting input of IC 4 - 12  is connected to the current sensing resistor Rs shown in  FIG. 4A . The inverting input of comparator IC 4 - 7  is connected to the output of the Current Source (V 0 ). The resistors R 4 - 5 , R 4 - 6  and capacitor C 4 - 8  are connected in series, where the common point of resistor R 4 - 6  and capacitor C 4 - 8  is connected to the inverting input of IC 4 - 8 . Because of rectifier D 4 - 3  connected to the common point of resistor R 4 - 5  and R 4 - 6 , the voltage on the inverting input is effected by the output voltage V 0  if it is lower then approximately 11V. The corresponding signal wave forms are shown in  FIG. 4D . If the output current increases above a certain level, than V 46 =0, and the monostable circuit of Timer unit will be triggered implementing peak current limitation. When the output voltage V 0 , depending on the load impedance, decreases bellow approximately 11V, the output V 48  goes to 1 and Current Source switches off, implementing short circuit protection. The main advantage of this solution that the actual short circuit operation exists only for a short time and the ballast is switched off until the short circuit condition exists (nearly zero output impedance).       

     FIG. 4F  and  FIG. 4G  show a detailed illustration of the transition process from high frequency to low frequency operation and the short circuit protection. As it was previously described the Current Limiter unit switches off both the lower switches of the inverter and the Current Source for a certain predetermined time if the current reaches a certain level, for instance  20 A. This way the maximum peak current in the MOSFET&#39;s can be limited to a safe level, even at increased temperature. 
   Thus, while preferred embodiments of the present invention have been shown and described in detail, it is to be understood that such adaptations and modifications as occur to those skilled in the art may be employed without departing from the spirit and scope of the invention, as set forth in the claims.