Patent Publication Number: US-10320278-B2

Title: Semiconductor device having a decreased switching loss

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation application, under 35 U.S.C § 111(a), of International Patent Application No. PCT/JP2016/068388, filed Jun. 21, 2016, and based upon the benefit of foreign priority from Japanese Patent Application No. 2015-169709, filed Aug. 28, 2015, the contents of which are incorporated herein by reference. 
    
    
     TECHNICAL FIELD 
     The present invention relates to a semiconductor device which has a first semiconductor element and a second semiconductor element connected in series and which is applied to a power conversion apparatus and so forth. 
     BACKGROUND ART 
     Conventionally, an inverter is applied to an electric motor, a vacuum cleaner, an air-conditioner, a welding machine and so forth as the power conversion apparatus. The semiconductor device in which the first semiconductor element and second semiconductor element are connected in series is used in the inverter and a power factor improvement circuit, a brake circuit and so forth which are peripheral circuits thereof. 
     In the inverter that a plurality of lines of the series-connected first semiconductor element and second semiconductor elements are connected in parallel, in general, it is intended to use switching semiconductor elements of the same kind such as a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), an IGBT (Insulated Gate Bipolar Transistor) and so forth as the first semiconductor elements which constitute an upper arm and the second semiconductor elements which constitute a lower arm. 
     Then, for example, in an inverter control circuit used in the vacuum cleaner, as a lower-arm-side switching element which constitutes the inverter control circuit, a switching element (for example, a wide band gap semiconductor element which uses gallium nitride (GaN), silicon carbide (SiC), diamond and so forth) which makes switching of a speed which is higher than that of an upper-arm-side switching element (for example, the IGBT) possible is applied (for example, see Patent Literature 1). In this Patent Literature 1, inverter control of a two-phase modulation system which is so made as to turn the upper-arm switching element on and turn the lower-arm switching element off in turn at intervals of 2π/3 at each phase voltage of three phase voltages to be applied to the motor thereby to periodically fix each phase voltage is performed. 
     CITATION LIST 
     Patent Literature 
     PTL 1: JP 2012-249488 A 
     SUMMARY OF INVENTION 
     Technical Problem 
     Incidentally, in the conventional example described in Patent Literature 1, a switching element which is faster in switching speed than a switching element on the upper arm side is used on the lower arm side and further the inverter control of the two-phase modulation system is performed so as to reduce a variation in switching loss (heat generation) between switching elements of the upper and lower arms. 
     However, there is such a problem that it is impossible to sufficiently improve the switching loss when turning-on of the lower-arm switching element simply by making the switching speed of the lower-arm switching element faster than the switching speed of the upper-arm switching element as in the above-mentioned conventional example. 
     Accordingly, the present invention is made focusing on the problem of the above-mentioned conventional example and to provide a semiconductor device which is capable of reducing the total loss of the upper and lower arms by reducing switching losses when turning-on of the switching elements is set as an object thereof. 
     Solution to Problem 
     In order to attain the above-mentioned object, one aspect of the semiconductor device according to the present invention has a first semiconductor element and a second semiconductor element connected in series, in which the first semiconductor element includes a low switching loss semiconductor element having a switching loss which is smaller than a switching loss of the second semiconductor element and the second semiconductor element includes a low conduction loss semiconductor element having a conduction loss which is smaller than a conduction loss of the first semiconductor element. 
     Advantageous Effects of Invention 
     According to one aspect of the present invention, it is possible to reduce the switching losses when turning-on of the switching elements and thereby to reduce the total loss of the upper and lower arms. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a circuit diagram illustrating a first embodiment of a semiconductor device according to the present invention; 
         FIGS. 2A and 2B  are waveform diagrams illustrating a three-phase sine-wave control waveform and a lower-side-stuck two-phase modulation control waveform; 
         FIGS. 3A to 3I  are signal waveform diagrams of respective phases of a lower-side-stuck two-phase modulation system in the first embodiment; 
         FIGS. 4A and 4B  are explanatory diagrams illustrating a W-phase powder supply path in  FIG. 2B .  FIG. 4A  illustrates the power supply path in a conduction section T 1  and  FIG. 4B  illustrates the power supply path in a section T 2 ; 
         FIG. 5  is a circuit diagram of an intelligent power module which configures a U-phase arm in the first embodiment; 
         FIGS. 6A to 6D  are signal waveform diagrams illustrating a collector-to-emitter voltage and a collector current when turning-on and turning-off of the upper arm and the lower arm in  FIG. 5 ; 
         FIGS. 7A to 7B  are diagrams expressing simulation results of respective losses of one switching arm section, in which (a) is a graph expressing a loss of a comparative example and (b) is a graph of a loss of a second embodiment; 
         FIG. 8  is a circuit diagram illustrating the second embodiment of the semiconductor device according to the present invention; 
         FIG. 9  is a circuit diagram illustrating an arm for one phase of the comparative example corresponding to the second embodiment; 
         FIGS. 10A and 10B  are waveform diagrams of the collector-to-emitter voltage and the collector current when turning-on of the upper arm in  FIG. 9  and a waveform diagram of a reflux current and a reverse recovery voltage in reverse recovery of a lower-arm freewheeling diode; 
         FIGS. 11A and 11B  are diagrams expressing simulation results of respective losses of one switching arm section.  FIG. 11A  is a graph expressing the loss of the comparative example and  11 B is a graph of the loss of the second embodiment; 
         FIG. 12  is a circuit diagram illustrating a third embodiment of the semiconductor device according to the present invention; and 
         FIG. 13  is a circuit diagram illustrating an altered example of the semiconductor device according to the present invention. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Next, one embodiment of the present invention will be described with reference to the drawings. In the following description of the drawings, the same or similar symbols are assigned to the same or similar parts. 
     In addition, the embodiments which will be indicated below are the ones which exemplify a device and a method for embodying the technical idea of the present invention and the technical idea of the present invention does not specify materials, shapes, structures, arrangements and so forth of constitutional components to the ones mentioned below. It is possible for the technical idea of the present invention to add various alterations within a technical range that the claims described in the scope of patent claims define. 
     In the following, a driving device of a semiconductor element according to one embodiment of the present invention will be described with reference to the drawings. In the present embodiment, description will be made by taking a voltage drive type semiconductor element as an example of the semiconductor element and taking a gate driving device of the semiconductor element as an example of the driving device of the semiconductor element. 
     First, a power conversion apparatus  10  which is equipped with a semiconductor device according to the present embodiment will be described by using  FIG. 1 . 
     As illustrated in  FIG. 1 , the power conversion apparatus  10  is equipped with a full-wave rectification circuit  12  which converts three-phase AC power which is input from a three-phase AC power source  11  into DC power, a smoothing capacitor  13  which smooths out the DC power which is output through a positive electrode line Lp and a negative electrode line Ln of the full-wave rectification circuit  12 , a brake circuit  14  which is connected in parallel with the smoothing capacitor  13 , an inverter unit  15  which is connected in parallel with the brake circuit  14  and which drives a three-phase motor  17  as a load, and a control unit  16  which controls the brake circuit  14  and the inverter unit  15 . 
     The full-wave rectification circuit  12  includes a full-bridge circuit in which series circuits  12 A,  12 B and  12 C in which two diodes are connected in series are connected in parallel between the positive electrode line Lp and the negative electrode line Ln. Each phase power of the three-phase AC power source  11  is supplied to a connection point between the diodes of each of the series circuits  12 A,  12   b  and  12 C and each phase power is subjected to full-wave rectification by each diode and the DC power is output from between the positive electrode line Lp and the negative electrode line Ln. 
     The brake circuit  14  is made so as to consume a regenerative current through a resistor for protection from an overvoltage which is applied to the inverter unit  15  when subjecting the three-phase motor  17  to regenerative braking. The brake circuit  14  includes a surge absorbing diode  14   a , a switching semiconductor element  14   b , and an externally attached resistor  14   c  connected in parallel with the diode  14   a.    
     Then, a cathode of the diode  14   a  and one end of the resistor  14   c  are connected to the positive electrode line Lp and a connection point of an anode of the diode  14   a  with the other end of the resistor  14   c  is connected to a collector of an insulated gate bipolar transistor (in the following, referred to as an IGBT) which works as the switching semiconductor element  14   b . An emitter of the IGBT is connected to the negative electrode line Ln and a gate thereof is connected to the control unit  16 . 
     The inverter unit  15  is equipped with a U-phase switching arm section  15 U, a V-phase switching arm section  15 V and a W-phase switching arm section  15 W which are connected in parallel between the positive electrode line Lp and the negative electrode line Ln. 
     The U-phase switching arm section  15 U is, an upper arm part which is connected to the positive electrode line Lp is constituted a first semiconductor element Su which includes an N-channel type MOSFET (Metal-Oxide-Semiconductor Field-Effect transistor) which is smaller in switching loss than the IGBT and a freewheeling diode Du connected in anti-parallel with this first semiconductor element Su. 
     The U-phase switching arm section  15 U is, a lower arm part connected to the negative electrode line Ln is constituted by a second semiconductor element Sx which includes the IGBT which is smaller in conduction loss than the MOSFET and a freewheeling diode Dx connected in anti-parallel with this second semiconductor element Sx. 
     Then, a drain of the MOSFET which includes the first semiconductor element Su is connected to the positive electrode line Lp, a source thereof is connected to a collector of the IGBT which includes the second semiconductor element Sx, and an emitter of the IGBT which includes the second semiconductor element Sx is connected to the negative electrode line Ln. 
     The V-phase switching arm section  15 V is, an upper arm part which is connected to the positive electrode line Lp is constituted by a first semiconductor element Sv which includes the MOSFET and a freewheeling diode Dv which is connected in anti-parallel with this first semiconductor element Sv, and a lower arm part which is connected to the negative electrode line Ln is constituted by a second semiconductor element Sy which includes the IGBT and a freewheeling diode Dy connected in anti-parallel with this second semiconductor element Sy. 
     The W-phase switching arm section  15 W is, an upper arm part which is connected to the positive electrode line Lp is constituted by a first semiconductor element Sw which includes the MOSFET and a freewheeling diode Dw which is connected in anti-parallel with this first semiconductor element Sw, and a lower arm part which is connected to the negative electrode line Ln is constituted by a second semiconductor element Sz which includes the IGBT and a freewheeling diode Dz which is connected in anti-parallel with this second semiconductor element Sz. 
     Relations of connection between the first semiconductor elements Sv, Sw and the second semiconductor elements Sy, Sz of these V-phase switching arm section  15 V and W-phase switching arm section  15 W are made the same as the aforementioned relation of connection between them of the U-phase switching arm section  15 U. 
     Then, gates of the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz of the switching arm sections  15 U- 15 W of respective phases are connected to the control unit  16 . 
     A U-phase output, a V-phase output and a W-phase output which are output from connection points between the first semiconductor elements Su, Sv, Sw and the second semiconductor elements Sx, Sy, Sz of the switching arm sections  15 U,  15 V and  15 W of the respective phases are respectively output to a U-phase winding wire, a V-phase winding wire and a W-phase winding wire of the three-phase motor  17  as the load. 
     A U-phase current detection value, a V-phase current detection value and a W-phase current detection value are input into the control unit  16  from current detectors  19   u ,  19   v  and  19   w  which detect output currents of output lines Lu, Lv and Lw of the inverter unit  15 . In addition, a voltage detection value is input into the control unit  16  from a voltage detector  20  which detects a terminal voltage between both ends of the smoothing capacitor  13 . 
     The control unit  16  controls a gate of the IGBT which serves as the fourth semiconductor element of the brake circuit  14  such that the terminal voltage does not become an overvoltage due to regenerative power which is input from the three-phase motor  17  when subjected to regenerative braking on the basis of the terminal voltage which is input from the voltage detector  20 . 
     In addition, the control unit  16  controls a first semiconductor element  15   a  and a second semiconductor element  15   b  of the U-phase switching arm section  15 U with lower-side-stuck two-phase modulation on the basis of the current detection values which are input from the current detectors  19   u - 19   w  and a not illustrated current command value. 
     Here, the lower-side-stuck two-phase modulation control is a method of expressing a three-phase alternating current by seeing other two phases always from a phase of a minimum voltage by a three-phase AC voltage. Consequently, it results in expression of the three-phase alternating current by two-phase alternating current. That is, it means that in a section A of the three-phase alternating current illustrated in  FIG. 2A , the W phase is at the lowest voltage and the three-phase alternating current is expressed by the voltages of the U phase and V phase viewed from the W phase. Although in a balanced three-phase current, it is defined by a three-phase AC waveform which is different only in phase, it indicates that, in reality, the balanced three-phase alternating current can be expressed by two AC waveforms. 
     When this two-phase modulation system is adopted, the AC waveform becomes a bottom-shaped waveform that any one of phases has stuck to the zero potential at intervals of 120 degrees as illustrated in  FIG. 2B . Drive waveforms of the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz of the switching arm sections  15 U- 15 W of the respective phases by the control unit  16  in order to obtain this bottom-shaped waveform are illustrated in  FIGS. 3A to 3I . 
     The operation of this control unit  16  will be described in regard to sections T 1  and T 2  that the W-phase sticks to the zero potential in  FIG. 2B . 
     In the section T 1 , as illustrated in  FIG. 3B , the first semiconductor element Su of the U-phase switching arm section  15 U which serves as an upper arm is driven with pulse width modulation (PWM) in a Hi duty ratio which is large in duty ratio (a pulse width for instructing ON of the semiconductor element is wide). In addition, as illustrated in  FIG. 3C , the first semiconductor element Sv of the V-phase switching arm section  15 V is driven with pulse width modulation (PWM) in an intermediate duty ratio. Further, as illustrated in  FIG. 3D , the first semiconductor element Sw of the W-phase switching arm section  15 W is controlled to an OFF state. 
     On the other hand, as illustrated in  FIG. 3E , the second semiconductor element Sx of the U-phase switching arm section  15 U which serves as a lower arm enters the OFF state and a reflux current flows to the freewheeling diode Dx. In addition, as illustrated in  FIG. 3F , the second semiconductor element Sy of the V-phase switching arm section  15 V is in the OFF state and the reflux current flows to the freewheeling diode Dy. Further, as illustrated in  FIG. 3G , the second semiconductor element Sz of the W-phase switching arm section  15 W is controlled to an always-ON state. 
     Therefore, a current path of the inverter unit  15  and the three-phase motor  17  in the section T 1  flows as indicated by a bold solid line L 1  in  FIG. 4A . That is, a motor current flows from the positive electrode line Lp through the first semiconductor element Su of the U-phase switching arm section  15 U, through a U-phase winding wire of the three-phase motor  17  and respectively through a V-phase winding wire and a W-phase winding wire from a neutral point and through the second semiconductor element Sy or the freewheeling diode Dy of the V-phase switching arm section  15 V (the current flows to the second semiconductor element Sy as it is or flows in a form that the reflux current of the freewheeling diode Dy is reduced depending on the orientation of the current which actually flows to the V-phase. The same shall apply hereinafter) and the second semiconductor element Sz of the W-phase switching arm section  15 W to the negative electrode line Ln. 
     On the other hand, the reflux current from the three-phase motor  17  flows as indicated by a bold solid line L 2 . That is, the reflux current from the V-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Dv of the upper arm part of the V-phase switching arm section  15 V—the positive electrode line Lp—the first semiconductor element Su of the U-phase switching arm section  15 U toward the U-phase winding wire of the three-phase motor  17 . In addition, the reflux current from the W-phase winding wire of the three-phase motor  17  flows through the second semiconductor element Sz of the W-phase switching arm section  15 W—the negative electrode line Ln—the freewheeling diode Dx of the lower arm part of the U-phase switching arm section  15 U toward the U-phase winding wire of the three-phase motor  17 . 
     Further, when all the semiconductor elements are in the OFF states, the reflux current from the three-phase motor  17  flows as indicated by a fine broken line L 3  in  FIG. 4A . That is, the reflux current from the V-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Dv of the upper arm part of the V-phase switching arm section  15 V—the positive electrode line Lp—the smoothing capacitor  13 —the negative electrode line—the freewheeling diode Dx of the lower arm part of the U-phase switching arm section  15 U to the U-phase winding wire of the three-phase motor  17 . In addition, the reflux current from the W-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Dw of the upper arm part of the W-phase switching arm section  15 W—the positive electrode line Lp—the smoothing capacitor  13 —the negative electrode line—the freewheeling diode Dx of the lower arm part of the U-phase switching arm section  15 U to the U-phase winding wire of the three-phase motor  17 . 
     On the other hand, in the section T 2 , the inverter unit  15  is controlled by the control unit  16  as illustrated in  FIGS. 3A to 3C . That is, as illustrated in  FIG. 3B , the first semiconductor element Su of the U-phase switching arm section  15 U is controlled to the OFF state. In addition, as illustrated in  FIG. 3C , the first semiconductor element Sv of the V-phase switching arm section  15 V is controlled with pulse width modulation in the Hi duty ratio of a comparatively wide pulse width. Further, as illustrated in  FIG. 3D , the first semiconductor element Sw of the W-phase switching arm section  15 W is controlled to the OFF state. 
     Simultaneously therewith, as illustrated in  FIG. 3E , the second semiconductor element Sx of the U-phase switching arm section  15 U is controlled with pulse width modulation (PWM) in the intermediate duty ratio. In addition, as illustrated in  FIG. 3F , the second semiconductor element Sy of the V-phase switching arm section  15 V is controlled to the OFF state and the reflux current flows to the freewheeling diode Dy. Further, as illustrated in  FIG. 3G , the second semiconductor element Sz of the W-phase switching arm section  15 W is controlled to the always-ON state. 
     Therefore, the current path of the inverter unit  15  and the three-phase motor  17  in the section T 2  flows as indicated by a bold solid line L 4  in  FIG. 4B . That is, the motor current flows from the positive electrode line Lp through the first semiconductor element Sv of the V-phase switching arm section  15 V, through the V-phase winding wire of the three-phase motor  17  and respectively through the U-phase winding wire/the freewheeling diode Dx and the W-phase winding wire from the neutral point, and through the second semiconductor element Sx of the U-phase switching arm section  15 U and the second semiconductor element Sz of the W-phase switching arm section  15 W to the negative electrode line Ln. 
     On the other hand, the reflux current from the three-phase motor  17  flows as indicated by a solid solid line L 5 . That is, the reflux current from the U-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Du of the upper arm part of the U-phase switching arm section  15 U—the positive electrode line Lp—the first semiconductor element Sv of the V-phase switching arm section  15 V toward the V-phase winding wire of the three-phase motor  17 . In addition, the reflux current from the W-phase winding wire of the three-phase motor  17  flows through the second semiconductor element Sz of the W-phase switching arm section  15 W—the negative electrode line Ln—the freewheeling diode Dx of the lower arm part of the U-phase switching arm section  15 U toward the V-phase winding wire of the three-phase motor  17 . 
     Further, when all the semiconductor elements are in the OFF states, the reflux current from the three-phase motor  17  flows as indicated by the fine broken line L 6  in  FIG. 4B . That is, the reflux current from the U-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Du of the upper arm part of the U-phase switching arm section  15 U—the positive electrode line Lp—the smoothing capacitor  13 —the negative electrode line—the freewheeling diode Dy of the lower arm part of the V-phase switching arm section  15 V to the V-phase winding wire of the three-phase motor  17 . In addition, the reflux current from the W-phase winding wire of the three-phase motor  17  flows through the freewheeling diode Dw of the upper arm part of the W-phase switching arm section  15 W—the positive electrode line Lp—the smoothing capacitor  13 —the negative electrode line—the freewheeling diode Dx of the lower arm part of the U-phase switching arm section  15 U to the V-phase winding wire of the three-phase motor  17 . 
     Accordingly, as illustrated in  FIG. 2B , a U-phase two-phase modulated waveform increases from the start of the section T 1 , takes a peak value and thereafter decreases, keeps decreasing in the section T 2  and lowers down to the lowest potential (a potential which is noted as −0.200 in the drawing. In the following, it is also referred to as the zero potential). On the contrary, as illustrated in  FIG. 2B , a V-phase two-phase modulated waveform begins increasing from the zero potential at the start of the section T 1 , keeps increasing also in the section T 2 , takes the peak value and thereafter decreases. Further, as illustrated in  FIG. 2B , a W-phase two-phase modulated waveform maintains the zero potential between the section T 1  and the section T 2 . 
     Eventually, as illustrated in  FIG. 2B , the U-phase two-phase modulated waveform, the V-phase two-phase modulated waveform and the W-phase two-phase modulated waveform respectively become the same waveform which has the bottom-shaped waveform on the upper side and maintains the zero potential through 120 degrees at intervals of 240 degrees (2π/3) in electrical angle and have a phase difference of 120 degrees among the respective phases. 
     For example, a U-phase voltage has a trapezoidal waveform as illustrated in  FIG. 3H  and a U-phase current becomes a state which is approximate to a sine wave as illustrated in FIG.  3 I by controlling the first semiconductor elements Su-Sw and the second semiconductor element Sx-Sz which configure the inverter unit  15  with two-phase modulation in this way. Incidentally,  FIGS. 3H, 3I  correspond to cases where the above-mentioned Hi duty ratio and intermediate ratio are fixed. 
     Since, in this two-phase modulation control, any one of the second semiconductor elements Sx-Sz of any one of the switching arm sections  15 U- 15 W is always maintained in the always-ON state with no switching, it is possible to reduce a switching loss in comparison with the case of performing three-phase sine-wave driving by that amount. In addition, the power source utilization factor is high and a maximum value of an inter-line voltage of the three-phase motor  17  which works as the load becomes a DC voltage Vdc which is output from the full-wave rectification circuit  12 . 
     In contrast to this, in the case of conventional three-phase sine-wave driving, the first semiconductor elements and the second semiconductor elements which constitute switching arm sections of three phases are always controlled with pulse width modulation (PWM). Therefore, in the three-phase sine-wave driving, the switching loss is much, the power source utilization factor is low and the maximum value of the inter-line voltage of the three-phase motor  17  which works as the load amounts to √3 Vdc/2=0.86 Vdc of the DC voltage Vdc which is output from the full-wave rectification circuit  12 . 
     It becomes possible to reduce the switching loss for the three-phase sine-wave control by controlling the inverter unit  15  with two-phase modulation in this way. 
     Incidentally, in a case of controlling the inverter unit  15  with two-phase modulation, as described above, there are lower-side-stuck two-phase modulation control that the second semiconductor elements Sx-Sz of the lower arm part are set to the always-ON states without making them perform the switching operations and the bottom-shaped waveform is formed on the upper side as described above and upper-side-stuck two-phase modulation control which is described in Patent Literature 1 and that the first semiconductor elements Su-Sw of the upper arm part are set to the always-ON states without making them perform the switching operations and the bottom-shaped waveform is formed on the lower side. 
     However, the lower-side-stuck two-phase modulation control which has been described in the above-mentioned embodiment is capable of more reducing the switching loss than the upper-side-stuck two-phase modulation control. 
     The reason therefor is as follows. In general, the inverter unit  15  is constituted as an intelligent power module (IPM) that three arms of the U-phase switching arm section  15 U, the V-phase switching arm section  15 V and the W-phase switching arm section  15 W are made as one module. 
     This intelligent power module is, a circuit configuration in a case where, for example, the U-phase switching arm section  15 U is extracted is as illustrated in  FIG. 5 . 
     That is, an intelligent power module  25  is equipped with the first semiconductor element Su and the freewheeling diode Du which constitute the upper arm part of the U-phase switching arm section  15 U which is connected in series between a positive electrode terminal P and a negative electrode terminal N and the second semiconductor element Sx and the freewheeling diode Dx which constitute the lower arm part thereof. A first gate drive circuit GDu 1  is connected to a gate of the first semiconductor element Su and a second gate drive circuit GDu 2  is connected to a gate of the second semiconductor element Sx. The first gate drive circuit GDu 1  and the second gate drive circuit GDu 2  are respectively connected to power source terminals Vcc H  and Vcc L  to which the positive electrode side of a DC control power source  26  is connected and are connected to a common ground terminal COM to which a connection point of the negative electrode side of the DC control power source  26  with the ground is connected and a controlled power source is applied thereto. 
     The first gate drive circuit GDu 1  is provided with a reference voltage terminal Vs to which an emitter of the first semiconductor element Su is connected and a reference voltage at this reference voltage terminal Vs becomes the reference of a gate drive signal of the first semiconductor element Su. 
     In addition, a parallel circuit of the DC power source Vdc with the smoothing capacitor C is connected between the positive electrode terminal P of the intelligent power module  25  and the ground and a shunt resistor Rs which detects the U-phase current is connected between the negative electrode terminal N and the ground. 
     The first gate drive circuit GDu 1  and the first semiconductor element Su are, the reference voltage terminal Vs is connected to the emitter of the first semiconductor element Su by internal wiring in this way. Therefore, an impedance between the emitter of the first semiconductor element Su and the reference voltage terminal Vs is limited only to an internal wiring impedance and therefore has a small value. 
     On the other hand, the second gate drive circuit GDu 2  and the second semiconductor element Sx are, an emitter of the second semiconductor element Sx is connected to the common ground terminal COM via the shunt resistor Rs. Therefore, they are influenced by a common impedance in a large impedance including those of the shunt resistor Rs and external wiring. 
     Accordingly, as illustrated in  FIG. 6A , a switching characteristic when turning-on of the first semiconductor element Su is, a collector current Ic becomes a sharp waveform which is steep in ±di/dt and a switching loss Eon when turning-on becomes as relatively little as 0.24 mJ as indicated by a characteristic line L 21 . 
     In contrast, as illustrated in  FIG. 6B , the switching characteristic when turning-on of the second semiconductor element Sx is, ±di/dt becomes duller than that of the first semiconductor element Su by being influenced by the wiring impedance of the shunt resistor Rs for current detection and thereby the switching loss Eon when turning-on becomes 0.38 mJ and becomes more worse than that of the first semiconductor element Su. 
     In regard to the switching losses Eoff when turning-off of the first semiconductor element Su and the second semiconductor element Sx, as illustrated in  FIG. 6C  and  FIG. 6D , the first semiconductor element Su is Eoff=0.12 mJ and the second semiconductor element Sx is Eoff=0.14 mJ, and no large difference occurs between them. 
     In a case where the intelligent power module  25  is configured in this way, the switching loss of the second semiconductor element Sx which serves as the lower-arm-side one becomes worse more than the switching loss of the first semiconductor element Su which serves as the upper-arm-side one. 
     Accordingly, it becomes possible to provide a section that the second semiconductor elements Sx-Sz are controlled to the always-ON states by performing lower-side-stuck two-phase modulation control on the inverter unit  15  by the control unit  16  as described above, and thereby it becomes possible to reduce the switching losses of the second semiconductor elements Sx-Sz. 
     As a result, it becomes possible to compensate for worsening of the switching losses of the second semiconductor elements Sx-Sz in a case of constituting the intelligent power module  25 . Therefore, it becomes possible to reduce the total switching loss of the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz. In contrast, in the upper-side-stuck two-phase modulation control for forming the bottom-shaped waveform on the lower side in Patent Literature 1, it results in increasing the total switching loss inversely. 
     Further, in regard to a case of performing the above-mentioned lower-side-stuck two-phase modulation control on the inverter unit  15 , a result that a simulation has been made on a comparative example that the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz include the same IGBTs and the diodes are formed as the same freewheeling diodes is illustrated in  FIG. 7A . In a case of this comparative example, the conduction loss of the first semiconductor element Su on the upper arm side was Von=1.8 mJ, the switching loss thereof when turning on was ton=3.1 mJ, the switching loss thereof when turning-off was toff=1.4 mJ, the conduction loss of the freewheeling diode Du was Vf=0.14 mJ and the recovery loss thereof was trr=0.1 mJ. 
     On the other hand, the conduction loss of the second semiconductor element Sx on the lower arm side was Von=2.2 mJ, the switching loss thereof when turning-on was ton=0.7 mJ, the switching loss thereof when turning-off was toff=0.3 mJ, the conduction loss of the freewheeling diode Dx was Vf=0.6 mJ and the recovery loss thereof was trr=0.5 mJ. Then, the total loss was 10.9 mJ. 
     From the result of this simulation in  FIG. 7A , the switching loss when turning-on of the first semiconductor element Su is predominant on the upper arm part. The switching loss when turning-on of this first semiconductor element Su also depends on the characteristic of the freewheeling diodes of the lower arm part. 
     In contrast, on the lower arm part, the conduction loss of the second semiconductor element Sx is predominant. 
     Therefore, a result of simulation made in a case where the MOSFETs whose switching loss when turning-on is little in comparison with the IGBT are used as the first semiconductor elements Su-Sw and the IGBTs whose conduction loss is little in comparison with the MOSFET are used as the second semiconductor elements Sx-Sz as in the above-mentioned first embodiment is illustrated in  FIG. 7B . 
     As apparent from this  FIG. 7B , by using the MOSFETs as the first semiconductor elements and using the IGBTs as the second semiconductor elements, the conduction loss of the first semiconductor elements Su-Sw becomes Von=0.7 mJ, the switching loss thereof when turning-on becomes ton=2.7 mJ, the switching loss thereof when turning-off becomes toff=0.2 mJ, the conduction loss of the freewheeling diodes Du-Dw becomes Vf=0.09 mJ and the recovery loss thereof becomes trr=0.1 mJ and they are greatly reduced relative to those of the comparative example. 
     On the other hand, the conduction loss of the second semiconductor elements Sx-Sz is Von=2.2 mJ with no change from that of the comparative example and the switching loss thereof when turning-on reaches ton=1.5 mJ which is increased to about two times that of the comparative example. In addition, the switching loss thereof when turning-off is toff=0.3 mJ and the conduction loss of the freewheeling diodes Dx-Dz is Vf=0.6 mJ with no change from those of the comparative example and the recovery loss thereof amounts to trr=0.6 mJ which is slightly increased relative to that of the comparative example. 
     Then, the total loss becomes 9.0 mJ and it became possible to improve the total loss by about 15% in comparison with that of the comparative example in  FIG. 7A . 
     Incidentally, although in the above-mentioned first embodiment, a case where the Si-MOSFETs are used as the first semiconductor elements Su-Sw is described, they are not limited to them and wide bandgap semiconductor elements made of SiC, GaN, diamond and so forth which are less than them in switching loss when turning-on may be applied as the first semiconductor elements Su-Sw. In this case, it becomes possible to more reduce the total loss. 
     Next, a second embodiment of the semiconductor device according to the present invention will be described accompanied by  FIG. 8 . 
     In this second embodiment, in place of the case where the semiconductor elements which are little in switching loss are used as the first semiconductor elements Su-Sw and the semiconductor elements which are little in conduction loss are used as the second semiconductor elements Sx-Sz as in the aforementioned first embodiment, common semiconductor switching elements are used as the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz and only the freewheeling diodes of the lower arm part are changed to diodes of the switching loss which is smaller than the switching loss of those of the upper arm part. 
     That is, in the second embodiment, as illustrated in  FIG. 8 , the IGBTs which are little in conduction loss are used as the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz of the inverter unit  15  and Si-freewheeling diodes which are large in switching loss are used as the freewheeling diodes Du-Dw of the upper arm part. 
     In addition, SiC-Schottky diodes which are wide bandgap diodes that the switching loss (the recovery loss due to a reverse recovery current) is less than the switching loss of the Si-freewheeling diodes are used as the freewheeling diodes Dx-Dz of the lower arm part. Other configurations have configurations which are the same as those of the aforementioned first embodiment and the same symbols are assigned to parts corresponding to those in  FIG. 1  and detailed description thereof is omitted. 
     This second embodiment is the one made by focusing on the switching losses of the freewheeling diodes Dx-Dz of the lower arm parts of the U-phase switching arm section  15 U, the V-phase switching arm section  15 V and the W-phase switching arm section  15 W. 
     As described before in the first embodiment, the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz of the inverter unit  15  include the same semiconductor switching elements and the freewheeling diodes thereof also include the same Si-freewheeling diodes. In a case where the lower-side-stuck two-phase modulation control is performed on this inverter unit  15 , the switching loss when turning-on of the first semiconductor elements Su-Sw is predominant. 
     In this second embodiment, it is made so as to reduce the switching loss when turning-on of the first semiconductor elements Su-Sw by reducing the switching loss (the recovery losses) of the freewheeling diodes of the lower arm which faces the upper arm. 
     That is, the switching loss will be examined in regard to each of the U-phase switching arm section  15 U, the V-phase switching arm section  15 V and the W-phase switching arm section  15 W. 
     For example, in a case where the U-phase switching arm section  15 U is made representative, the switching characteristics when using the Si-freewheeling diodes which are large in switching loss as the freewheeling diode Du of the upper arm part and the freewheeling diode Dx of the lower arm part as illustrated in  FIG. 9  are as illustrated in  FIGS. 10A and 10B . 
     That is, a case where the first semiconductor element Su of the upper arm part is turned on from the OFF state and the freewheeling diode Dx of the lower arm part is brought from the ON state into the OFF state is considered. A main circuit current which flows from a connection point of the first semiconductor element Su with the second semiconductor element Sx to the U-phase winding wire of the three-phase motor  17  at this time is referred to as IC and a reflux current which flows through the freewheeling diode Dx of the lower arm part from a U-phase coil is referred to as IF. 
     In this case, it enters a state where a reverse bias voltage is applied to the freewheeling diode Dx of the lower arm part from a forward biased state where the reflux current IF according to the previous-time main circuit current IC flows thereto and the reflux current IF begins decreasing as illustrated in  FIG. 10B . 
     On the other hand, since the first semiconductor element Su of the upper arm part is turned on from the OFF state, the main circuit current IC begins increasing from zero as illustrated in  FIG. 10A  and a collector-to-emitter voltage VCE of the first semiconductor element Su begins decreasing as illustrated in  FIG. 10A . 
     At this time, since the freewheeling diode Dx of the lower arm part is in a state where the reverse bias voltage is applied thereto, a large reverse recovery current flows in a direction opposite to the diode in a short time after the reflux current has been reduced to zero. This reverse recovery current reaches a negative-side peak value and thereafter returns to zero. 
     On the other hand, in the first semiconductor element Su of the upper arm part, a reverse recovery peak current of the freewheeling diode Dx of the lower arm part is superimposed on the main circuit current IC, it largely jumps up and thereafter has an almost fixed value according to a gate voltage of the first semiconductor element Su as illustrated in  FIG. 10A . 
     Accordingly, it becomes possible to reduce the switching loss when turning-on of the first semiconductor element Su of the upper arm part by using, for example, the SiC-freewheeling diode which is the wide bandgap diode which is small in reverse recovery current and is smaller in switching loss (recovery loss) than the Si-freewheeling diode as the freewheeling diode Dx of the lower arm part. Incidentally, as the freewheeling diodes Dx-Dz, SiC-Schottky diodes and diodes in which a JBS (Junction Barrier Schottky) structure is applied may be used not limited to the SiC-freewheeling diodes and further diodes which are smaller than the Si-freewheeling diodes in switching loss (recovery loss) such as GaN-freewheeling diodes, diamond-freewheeling diodes and so forth may be used. 
     In regard to the freewheeling diodes Du-Dw of the upper arm part of each of the U-phase switching arm section  15 U, the V-phase switching arm section  15 V and the W-phase switching arm section  15 W, the Si-freewheeling diodes which are large in switching loss (recovery loss) are left as they are in this way. On the other hand, the SiC-freewheeling diodes which are smaller in switching loss (recovery loss) than the Si-freewheeling diodes are used for the freewheeling diodes Dx-Dz of the lower arm part which faces the upper arm. Thereby, it becomes possible to reduce the switching losses when turning-on of the first semiconductor elements Su-Sw and thereby to reduce the total loss. 
     A result of performing the simulation which is the same as that in the first embodiment on one switching arm section in this second embodiment is illustrated in  FIG. 11B . Incidentally,  FIG. 11A  illustrates the loss of a conventional example in which the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz are made as the IGBTs and the freewheeling diodes Du-Dw of the upper arm part and the freewheeling diodes Dx-Dz of the lower arm part are made as the Si-freewheeling diodes. 
     As apparent from this  FIG. 11B , according to the second embodiment, although the conduction loss of the first semiconductor elements Su-Sw is Von=1.8 mJ with no change from that of the conventional example, the switching loss when turning-on is greatly reduced to ton=1.4 mJ. In addition, although the switching loss when turning-off is toff=0.14 mJ with no change from that of the conventional example and the conduction loss of the freewheeling diode Du-Dw is Vf=0.14 mJ with no change from that of the conventional example, the recovery loss thereof is greatly reduced to trr=0.02 mJ. 
     In addition, in regard to the second semiconductor elements Sx-Sz, the conduction loss is Von=2.2 mJ, the switching loss when turning-on is ton=0.7 mJ, the switching loss when turning-off is toff=0.3 mJ, the conduction loss of the freewheeling diodes Dx-Dz is Vf=0.7 mJ and the recovery loss thereof is trr=0.5 mJ, and they have almost the same values as those of the conventional example. 
     Accordingly, the total loss becomes 9.2 mJ and it becomes possible to improve the total loss by about 12% in comparison with the conventional example in  FIG. 7A . 
     In this second embodiment, it becomes possible to reduce the switching losses when turning-on of the first semiconductor elements which become the upper arm part and thereby to reduce the total loss simply by using the IGBTs which are small in conduction loss for the first semiconductor elements Su-Sw and the second semiconductor elements Sx-Sz, by using the Si-freewheeling diodes which are large in switching loss (recovery loss) for the freewheeling diodes Du-Dw of the upper arm part, and by using the wide bandgap diodes which are small in switching loss (recovery loss) in comparison with the Si-freewheeling diodes only for the freewheeling diodes Dx-Dz of the lower arm part. 
     In addition, following the first embodiment, the first semiconductor elements Su-Sw may be made as N-channel type MOSFETs which are smaller in switching loss than the IGBT. 
     Incidentally, although in the above-mentioned embodiments, a case where the semiconductor device according to the present invention is applied to the inverter unit of the power conversion apparatus which drives the inductive load is described, it is not limited thereto and it is also possible to apply it to a controlling apparatus which makes a welding machine generate arc in place of the inductive load. 
     That is, as illustrated in  FIG. 12 , a controlling apparatus  30  for welding machine is equipped with a full-wave rectification circuit  32  which performs full-wave rectification on a commercial single-phase AC power source  31 , a power factor improvement circuit  33  which is connected to the positive electrode line Lp and the negative electrode line Ln through which the DC power is output from this full-wave rectification circuit  32 , a smoothing capacitor  34  which is connected between the positive electrode line Lp and the negative electrode line Ln on the output side of the power factor improvement circuit  33 , an inverter unit  35  which is connected in parallel with the smoothing capacitor  34  between the positive electrode line Lp and the negative electrode line Ln, a transformer  36  which is connected to the output side of the inverter unit  35 , and an output-side rectification circuit  37  which is connected to the secondary side of the transformer  36 . 
     Here, the full-wave rectification circuit  32  constitutes a single-phase full-bridge circuit by four diodes and converts commercial AC power into the DC power. 
     The power factor improvement circuit  33  has a configuration of a boost chopper which is equipped with a third semiconductor element  33   a  which is connected to the positive electrode line Lp, a fourth semiconductor element  33   b  which is connected between the input side of the third semiconductor element  33   a  and the negative electrode line Ln and a reactor  33   c  which is connected between a connection point of the third semiconductor element  33   a  with the fourth semiconductor element  33   b  and the positive electrode side of the full-wave rectification circuit  32 . 
     Here, the third semiconductor element  33   a  includes the wide bandgap semiconductor element such as the SiC-freewheeling diode, the SiC-Schottky diode, the GaN-diode, the diamond diode and so forth whose switching losses (recovery losses) are smaller than the switching loss (the recovery loss) of the Si-freewheeling diode. 
     The fourth semiconductor element  33   b  includes the IGBI which is the low conduction loss semiconductor element which is smaller in conduction loss than the MOSFET. 
     The inverter unit  35  is equipped with a first switching arm section  35 A in which a first semiconductor element Sa and a second semiconductor element Sc are connected in series between the positive electrode line Lp and the negative electrode line Ln and a second switching arm section  35 B in which a first semiconductor element Sb and a second semiconductor element Sd are connected in series between the positive electrode line Lp and the negative electrode line Ln and constitutes a single-phase full-bridge circuit by these first switching arm section  35 A and second switching arm section  35 B. 
     Freewheeling diodes Da and Db are connected in anti-parallel with the respective first semiconductor elements Sa and Sb and capacitors Ca and Cb are connected in parallel with these freewheeling diodes Da and Db. 
     Freewheeling diodes Dc and Dd are connected in anti-parallel with also the respective second semiconductor elements Sc and Sd and capacitors Cc and Cd are connected in parallel with these freewheeling diodes Dc and Dd. 
     The transformer  36  is connected to a connection point of the first semiconductor element Sa with the second semiconductor element Sc of the inverter unit  35  at one end of a primary side winding wire via a reactor  35   a  and is directly connected to a connection point between the first semiconductor element Sb and the second semiconductor element Sd at the other end thereof. The output-side rectification circuit  37  is connected to a secondary winding wire of the transformer  36 . 
     In the output-side rectification circuit  37 , anodes of diodes  37   a  and  37   b  are connected to both ends of the secondary side winding wire of the transformer  36 , cathodes of the diodes  37   a  and  37   b  are connected together and are connected to a positive electrode side output terminal tp, an intermediate tap of the secondary side winding wire of the transformer  36  is directly connected to a negative electrode side output terminal tn and a smoothing capacitor  37   c  is connected between the positive electrode side output terminal tp and the negative electrode side output terminal tn. Then, the positive electrode side output terminal tp and the negative electrode side output terminal to are connected to welding terminals of the welding machine and a workpiece. 
     According to this second embodiment, the commercial AC power which is input from the commercial single-phase AC power source  31  is subjected to full-wave rectification by the full-wave rectification circuit  32  and is converted into the DC power and this DC power is input into the power factor improvement circuit  33 . An output from this power factor improvement circuit  33  is stored into the smoothing capacitor  34 , is converted into a single-phase alternating current by the inverter unit  35 , thereafter is boosted by the transformer  36 , is converted into a direct current by the output-side rectification circuit  37  and is supplied to the welding machine. 
     Incidentally, in the power factor improvement circuit  33 , to store electric energy into the reactor  33   c  when the fourth semiconductor element  33   b  is in the ON state and to store the electric energy stored in the reactor  33   c  into the smoothing capacitor  34  via the third semiconductor element  33   a  while the fourth semiconductor element  33   b  is in the OFF state by preforming switching control on the fourth semiconductor element  33   b  by a not illustrated control unit are repeated. 
     At this time, since when the fourth semiconductor element  33   b  is in the OFF state, the electric energy which is stored in the reactor  33   c  flows to the smoothing capacitor  34  through the third semiconductor element  33   a  which includes the diode, the third semiconductor element  33   a  is in a forward-biased state. 
     When the fourth semiconductor element  33   b  enters a turn-on state from the OFF state in this forward-biased state, the anode side of the diode which configures the third semiconductor element  33   a  is connected to the negative electrode line Ln via the fourth semiconductor element  33   b . Therefore, it enters a state where a DC voltage which is stored in the smoothing capacitor  34  is applied to the cathode side of the diode as a reverse bias voltage. 
     Consequently, although the current which is flowing to the third semiconductor element  33   a  is gradually decreased at a predetermined decrease rate (−di/dt) and is decreased to zero once, thereafter, the large reverse recovery current flows in the direction opposite to the diode in a short time and this reverse recovery current is added to the fourth semiconductor element  33   b . Therefore, in a case where the Si-freewheeling diode which is large in switching loss (recovery loss) is used as the fourth semiconductor element  33   b , the reverse recovery current is superimposed on a collector current Ic of the fourth semiconductor element  33   b , the peak value of the collector current largely jumps up and the switching loss when turning-on of the fourth semiconductor element  33   b  is increased similarly to  FIG. 10A  of the aforementioned first embodiment. 
     However, in the above-mentioned second embodiment, the wide bandgap diode such as the SiC-freewheeling diode, the SiC-Schottky diode, the GaN-diode, the diamond diode and so forth which are sufficiently small in switching loss (recovery loss) in comparison with the Si-freewheeling diode is used as the third semiconductor element  33   a . Therefore, it becomes possible to greatly reduce the switching loss when turning-on of the fourth semiconductor element  33   b  and thereby to reduce the total loss similarly to the aforementioned first embodiment. 
     Incidentally, in the inverter unit  35 , it becomes possible to reduce the switching losses when turning-on of the first semiconductor elements Sa and Sb and thereby to reduce the total loss similarly to the aforementioned first embodiment by using the Si-freewheeling diode which is large in switching loss (recovery loss) for the freewheeling diodes Da and Db which become the upper arm part and by using the wide bandgap diodes of the switching loss (the recovery loss) which is smaller than the switching loss (the recovery loss) of the Si-freewheeling diode for the freewheeling diodes Dc and Dd which become the lower arm part. 
     It is possible to exhibit the same advantageous effect as that of the power factor improvement circuit in this case also by the brake circuit  14  in the first embodiment. 
     That is, as illustrated in  FIG. 13 , the diode  14   a  of the brake circuit  14  illustrated in  FIG. 1  and  FIG. 8  is made as a third semiconductor element  44   a , the switching semiconductor element  14   b  is made as a fourth semiconductor element  44   b  and the resistor  14   c  is made as a resistor  44   c.    
     The wide bandgap element which is less than a Si-pin diode in reverse recovery current, the wide bandgap element which is less than the Si-pin diode in switching loss (recovery loss) and is less than it in reverse recovery current such as the SiC-freewheeling diode, the GaN-freewheeling diode, a SiC-Schottky barrier diode and so forth is used for the third semiconductor element  44   a . A semiconductor element which is smaller than the MOSFET in conduction loss, for example, the IGBT is used for the fourth semiconductor element  44   b . In regard to other configurations, it has the same configurations as the first embodiment, the same symbols are assigned to the parts corresponding to those in  FIG. 1  in  FIG. 13  and detailed description thereof is omitted. 
     By constituting the brake circuit  14  in this way, it becomes possible to operate the three-phase motor  17  as a generator and thereby to perform regenerative braking by operating the inverter unit  15  as the full-wave rectification circuit by the control unit  16  when braking the three-phase motor  17  which becomes the load. 
     In this regenerative braking, the three-phase AC power which is output from the three-phase motor  17  is subjected to full-wave rectification by the inverter unit  15  and is stored into the smoothing capacitor  13 . At this time, the current flows through the resistor  44   c  and the fourth semiconductor element  44   b  and a current bypass line for the inverter unit  15  is formed by performing switching control on the fourth semiconductor element  44   b  of the brake circuit by the control unit  16 . In this occasion, rise in DC voltage Vdc applied to the inverter unit  15  is suppressed by consuming the power by the resistor  44   c . A surge voltage which is generated in this occasion is absorbed by the third semiconductor element  44   a.    
     In this brake circuit  44 , since the regenerative power is consumed by the resistor  44   c , the resistance value of the resistor  44   c  is set to a large value. Therefore, when it enters a regenerative braking state and the fourth semiconductor element  44   a  is made to perform the switching operation, the diode which configures the third semiconductor element  44   a  enters the reverse-biased state with the regenerative power. In a case where the Si-freewheeling diode is used as this third semiconductor element  44   a , similarly to the power factor improvement circuit  33  of the aforementioned welding machine control circuit, the reverse recovery current which flows through the third semiconductor element  44   a  becomes more to increase the switching loss when turning-on of the fourth semiconductor element  44   b.    
     However, it becomes possible to lower the switching loss (the recovery loss) and to suppress the peak value of the reverse recovery current by using the wide bandgap diode such as the SiC-Schottky diode and so forth as the fourth semiconductor element  44   b . Also in this case, similarly to the case in  FIG. 12 , it is possible to reduce the switching loss when turning-on of the fourth semiconductor element  44   b  and it is possible to reduce the total loss of the third semiconductor element  44   a  and the fourth semiconductor element  44   b.    
     Incidentally, although in the above-mentioned embodiments, the cases where the semiconductor device of the present invention is applied to the power conversion apparatus and the controlling apparatus for welding machine are described, it is not limited thereto and it is possible to apply the present invention to other apparatuses having series circuits for connecting the first semiconductor element and the second semiconductor element or the third semiconductor element and the fourth semiconductor element in series. 
     In addition, in regard to the one having the brake circuit in the above-mentioned embodiments, since the brake circuit is not required in all applications, the brake circuit may be omitted in an application which does not require the brake circuit. 
     REFERENCE SIGNS LIST 
       10  . . . power conversion apparatus,  11  . . . three-phase AC power source,  12  . . . full-wave rectification circuit,  13  . . . smoothing capacitor,  14  . . . brake circuit,  15  . . . inverter unit, Su-Sw . . . first semiconductor element, Sx-Sz . . . second semiconductor element, Du-Dw, Dx-Dz . . . freewheeling diode,  16  . . . control unit,  17  . . . three-phase motor,  25  . . . intelligent power module,  30  . . . controlling apparatus for welding machine,  31  . . . commercial single-phase AC power source,  32  . . . full-wave rectification circuit,  33  . . . power factor improvement circuit,  33   a  . . . third semiconductor element,  33   b  . . . fourth semiconductor element,  33   c  . . . reactor,  34  . . . smoothing capacitor,  35  . . . inverter unit, Sa, Sb . . . first semiconductor element, Sc, Sd . . . second semiconductor element, Da-Dd . . . freewheeling diode,  36  . . . transformer,  37  . . . output-side rectification circuit