Patent Publication Number: US-2005140402-A1

Title: Frequency converter

Description:
BACKGROUND OF THE INVENTION  
      This application claims the priority of Korean Patent Application No. 2003-96891, filed on Dec. 24, 2003, in the Korean Intellectual Property Office, the disclosure of which is incorporated herein in its entirety by reference.  
      1. Field of the Invention The present invention relates to a frequency converter and, more particularly, to a frequency converter that can remove a DC offset and a second order intermodulation distortion component.  
      2. Description of the Related Art A heterodyne architecture in a transmitting/receiving circuit on a wireless channel requires a plurality of separate components including a surface acoustic wave filter. Thus, it is difficult to produce a compact transceiver and reduce its consumption power.  
      A direct conversion receiver directly converts a received reference signal to a baseband signal without converting it to an intermediate frequency so the receiver can be integrated into one chip. However, in the direct conversion receiver, a local oscillator signal used for down conversion has a magnitude considerably larger than that of a received reference signal. Thus, it is difficult for the frequency converter in the direct conversion receiver to control the generation of DC offset caused by local oscillator signal self-mixing.  
      A DC offset is generated by the leakage of the local oscillator signal to the input of the frequency converter, which is mixed then with local oscillator signal and a DC offset is generated. The local oscillator signal can leak directly to the input port of the frequency converter, of in-directly by capacitive coupling, coupling through substrate and inductive coupling. The local oscillator signal can be leak through LNA because of finite reverse isolation of the LNA. The local oscillator leakage signal is amplified by the LNA and mixed with local oscillator signal. The DC offset saturates an automatic gain control (AGC) or a low pass filter, which is connected to the back end of the frequency converter, to cause a signal distortion and deteriorate the sensitivity of a receiver.  
      Furthermore, the frequency converter in the direct conversion receiver brings about second order intermodulation distortion. The second order intermodulation distortion is close proximity to a signal converted by the frequency converter. When an interference signal having a relatively large magnitude is input to the frequency converter, the magnitude of an output second order intermodulation distortion component is larger than that of a desired output signal component to result in a reduction in receiving sensitivity.  
      Accordingly, studies on the direct conversion in order to remove the DC offset and second order intermodulation distortion component have been performed. An even harmonic mixer with a local oscillation signal frequency which is one-half of the reference signal containing RF signal frequency is a representative direct conversion technique.  
       FIG. 1  shows the configuration of a conventional even harmonic mixer. Referring to  FIG. 1 , the even harmonic mixer includes a band pass filter  10 , a band rejection filter  20 , and an anti-parallel diode  30 . Specifically, the band pass filter  10  that amplifiers an input signal and a band rejection filter  20  that filters a noise of the input signal are located between an input signal port fi and an output signal port fo. The anti-parallel diode  30  is connected between the band pass filter  10  and band rejection filter  20 . The anti-parallel diode  30  includes first and second diodes  31  and  32  connected to each other. One end of the anti-parallel diode  30  is connected to an open circuit stub  40  and the other end is connected to a short circuit stub  50 .  
      The anti-parallel diode  30  has odd symmetrical characteristic and restricts an even-order distortion including self-mixing of a local oscillator signal LO according to the odd symmetrical characteristic. However, the magnitude of the local oscillator signal LO applied to control turning on/off of the diodes  31  and  32  is as large as more than 0 dBm so that it may produce a lot of leakage components in the even harmonic mixer.  
      To prevent the generation of the leakage components, another even harmonic mixer using transistors having a low DC offset has been proposed.  FIG. 2  shows the even harmonic mixer using the transistors. Referring to  FIG. 2 , the even harmonic mixer includes first and second circuits  60  and  70 . The first circuit  60  is constructed in a manner that a plurality of MOS transistors is connected in a differential amplifier form. Specifically, the first circuit  60  includes two sub differential circuits each of which has two MOS transistors connected in a differential amplifier form. Positive and negative local oscillating signal LO+ and LO−are respectively input to input ports of the MOS transistors constructing the sub differential circuits. The drains of the MOS transistors of each sub differential circuit are connected to each other. The first circuit  60  is connected to the second circuit  70  having MOS transistors connected in a differential amplifier form. Reference signals RF+ and RF− are applied to the MOS transistors of the second circuit  70 .  
      The even harmonic mixer outputs a mixed signal of an odd-order harmonics of the reference signal RF and an even-order harmonics of the local oscillator signal LO. That is, the even harmonic mixer can prevent the reference signal RF from being mixed with an odd-order harmonics of the local oscillator signal LO. When the reference signal RF and local oscillator signal LO are sin ω RF t and sin ω LO t respectively, an output voltage V BB (t) is represented as follows  
                       V   BB     ⁡     (   t   )       =       ⁢         V   1     ⁡     (   t   )       -       V   2     ⁡     (   t   )                     =       ⁢         (       4   ⁢     α   1       +     9   ⁢     α   3       +     35   ⁢     α   5         )     ⁢   sin   ⁢           ⁢     ω   RF     ⁢   t     -                     ⁢         (       3   ⁢     α   2       -       5   /   4     ⁢     α   5         )     ⁢   sin   ⁢           ⁢   3   ⁢     ω   RF     ⁢   t     +       5   /   4     ⁢   sin   ⁢           ⁢   5   ⁢           ⁢     ω   RF     ⁢   t     -                     ⁢         (       3   ⁢     α   3       +     5   ⁢     α   5         )     ⁢     sin   ⁡     (       ω   RF     ±     2   ⁢           ⁢     ω   LO         )       ⁢   t     +                     ⁢         5   /   4     ⁢     α   5     ⁢     sin   ⁡     (       ω   RF     ±     4   ⁢     ω   LO         )       ⁢   t     +                     ⁢         5   /   2     ⁢     α   5     ⁢     sin   ⁡     (       3   ⁢     ω   RF       ±     2   ⁢     ω   LO         )       ⁢   t     +   …                   [     Equation   ⁢           ⁢   1     ]             
 
       FIG. 3  shows the output spectrum of the even harmonic mixer, represented by Equation 1. Referring to Equation 1 and  FIG. 3 , the reference signal RF is downconverted by the second-order harmonic of the local oscillator signal LO to desired output signal ω RF −2ω LO  and mirror signal ω RF +2ω LO . And, the odd-order harmonics of the reference signal RF, ω RF  and 3ω RF , and mixed by even-order harmonics of the local oscillator signal LO, ω RF ±4ω LO  and 3ω RF ±2ω LO , appear in the output signal of the even harmonic mixer of  FIG. 2 . Therefore, the even harmonic mixer has high spurious response levels including even-order harmonics of the LO signal.  
      The output voltage V BB (t) when two closely spaced input tones ω a  and ω b  are input to the reference signal RF ports of the even harmonic mixer is represented as follows.  
                       V   BB     ⁡     (   t   )       =       ⁢           β   1     ⁡     (       sin   ⁢           ⁢     ω   a       +     sin   ⁢           ⁢     ω   b         )       ⁢   t     +                     ⁢         β   2     ⁢     {         (       sin   ⁢           ⁢   2   ⁢     ω   a       +     ω   b       )     ⁢   t     +       (       sin   ⁢           ⁢   2   ⁢     ω   b       -     ω   a       )     ⁢   t       }       +                     ⁢         β   3     ⁢   sin   ⁢           ⁢   2   ⁢     ω   LO     ⁢   t   ⁢           ⁢     sin   ⁡     (       ω   a     -     ω   b       )       ⁢   t     +                     ⁢         β   4     ⁢   sin   ⁢           ⁢   2   ⁢     ω   LO     ⁢   t   ⁢           ⁢     sin   ⁡     (       2   ⁢     ω   a       -     ω   b       )       ⁢   t     +                     ⁢         β   5     ⁢   sin   ⁢           ⁢   2   ⁢     ω   LO     ⁢   t   ⁢           ⁢     sin   ⁡     (       2   ⁢     ω   b       -     ω   a       )       ⁢   t     +   …                   [     Equation   ⁢           ⁢   2     ]             
 
      From Equation 2, it can be known that third-order intermodulation distortion products 2ω a −ω b −ω LO  and 2ω b −ω a −ω LO  related with circuit linearity are exist in the output signal while second-order intermodulation distortion products ω a −ω b  and ω b −ω a  are suppressed. Therefore, this mixer has low third-order intercept point.  
     SUMMARY OF THE INVENTION  
      The present invention provides a frequency converter that can remove a DC offset and suppress spurious responses and intermodulation distortion products.  
      According to an aspect of the present invention, there is provided a frequency converter that receives a reference signal and a local oscillator signal and outputs a frequency component corresponding to the sum or difference of a fundamental frequency of the reference signal and a second-order harmonic of the local oscillation signal. The frequency converter includes a reference signal input part including a pair of MOS transistors connected in a differential amplifier form, which have gates to which positive and negative reference signals having a differential phase difference therebetween are respectively input, and first, second, third and fourth frequency conversion parts each of which is connected to the reference signal input part and includes a pair of MOS transistors. Local oscillator signals having a differential phase difference therebetween are input to the gates of the MOS transistors of the first and second frequency conversion parts. Local oscillator signals, which have phases orthogonal to phases of the local oscillator signals input to the first and second frequency conversion parts, are input to the gates of the MOS transistors of the third and fourth frequency conversion parts.  
      The sources of the MOS transistors of the first and third frequency conversion parts are commonly connected to the drain of the MOS transistor of the reference signal input part, to which the positive reference signal is applied. The sources of the MOS transistors of the second and fourth frequency conversion parts are commonly connected to the drain of the MOS transistor of the reference signal input part, to which the negative reference signal is applied.  
      The drains of the MOS transistors of the first and fourth frequency conversion parts are commonly connected to a first output port BB − , and the drains of the MOS transistors of the second and third frequency conversion parts are commonly connected to a second output port BB + .  
      According to another aspect of the present invention, there is provided a frequency converter that receives a reference signal and a local oscillator signal and outputs a frequency component corresponding to the sum or difference of the reference signal and the even-order harmonics of the local oscillator signal. The frequency converter includes a reference signal input part including a pair of bipolar transistors connected in a differential amplifier form, which have bases to which positive and negative reference signals having a differential phase difference therebetween are respectively input, and first, second, third and fourth frequency conversion parts each of which is connected to the reference signal input part and includes a pair of bipolar transistors connected in a differential amplifier form. Local oscillator signals having a differential phase difference therebetween are input to the bases of the bipolar transistors of the first and second frequency conversion parts. Local oscillator signals, which have phases orthogonal to phases of the local oscillator signals input to the first and second frequency conversion parts, are input to the bases of the bipolar transistors of the third and fourth frequency conversion parts. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
      The above and other features and advantages of the present invention will become more apparent by describing in detail exemplary embodiments thereof with reference to the attached drawings in which:  
       FIG. 1  shows the configuration of a conventional even harmonic mixer employing diodes;  
       FIG. 2  is a circuit diagram of a conventional even harmonic mixer employing transistors;  
       FIG. 3  shows output spectrum characteristic of the even harmonic mixer of  FIG. 1  and  FIG. 2 ;  
       FIG. 4  is a circuit diagram of a frequency converter according to an embodiment of the present invention;  
       FIG. 5  shows the output spectrum of an even harmonic mixer according to an embodiment of the present invention; and  
       FIG. 6  is a circuit diagram of a frequency converter according to another embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION  
      The present invention will now be described more fully with reference to the accompanying drawings, in which exemplary embodiments of the invention are shown. The invention may, however, be embodied in many different forms and should not be construed as being limited to the embodiments set forth herein; rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the concept of the invention to those skilled in the art. Throughout the drawings, like reference numerals refer to like elements.  
       FIG. 4  is a circuit diagram of a frequency converter according to an embodiment of the present invention. Referring to  FIG. 4 , the frequency converter includes a reference signal input part  110 , first, second, third and fourth frequency conversion parts  120 ,  130 ,  140  and  150 .  
      The reference signal input part  110  includes a pair of first and second MOS transistors Tra and Trb connected in a differential amplifier form. Reference signals RF+ and RF− having a differential phase difference therebetween are input to the gates of the first and second MOS transistors Tra and Trb. The sources of the MOS transistors Tra and Trb are coupled to each other and connected to a current source I.  
      The first frequency conversion part  120  is connected to the drain of the first MOS transistor Tra of the reference signal input part  110 . The first frequency conversion part  120  includes a pair of first and second MOS transistors Tr 1   a  and Tr 1   b  connected in a differential amplifier form. A local oscillator signal LO 0  is input to the gate of the first MOS transistor Tr 1   a  of the first frequency conversion part  120  and a local oscillator signal LO 180  having a differential phase difference from the local oscillator signal LO 0  is input to the gate of the second MOS transistor Tr 1   b . The sources of the first and second MOS transistors Tr 1   a  and Tr 1   b  of the first frequency conversion part  120  are coupled to each other and connected to the reference signal input part  110 . The drains of the first and second MOS transistors Tr 1   a  and Tr 1   b  are connected to a first output port BB − .  
      The second frequency conversion part  130  is connected to the drain of the second MOS transistor Trb of the reference signal input part  110 . The second frequency conversion part  130  includes a pair of first and second MOS transistors Tr 2   a  and Tr 2   b  connected in a differential amplifier form. The local oscillator signal LO 180  is input to the gate of the first MOS transistor Tr 2   a  of the second frequency conversion part  130  and the local oscillator signal LO 0  having a differential phase difference from the signal LO 180  is input to the gate of the second MOS transistor Tr 2   b . The sources of the first and second MOS transistors Tr 2   a  and Tr 2   b  of the second frequency conversion part  130  are coupled to each other and connected to the reference signal input part  110 . The drains of the first and second MOS transistors Tr 2   a  and Tr 2   b  are connected to a second output port BB ± .  
      The third frequency conversion part  140  is connected to the drain of the first MOS transistor Tra of the reference signal input part  110 . The third frequency conversion part  140  includes a pair of first and second MOS transistors Tr 3   a  and Tr 3   b  connected in a differential amplifier form. A local oscillating signal LO 90  is input to the gate of the first MOS transistor Tr 3   a  of the third frequency conversion part  140  and the local oscillator signal LO 270  having a differential phase difference from the signal LO 90  is input to the gate of the second MOS transistor Tr 3   b . The sources of the first and second MOS transistors Tr 3   a  and Tr 3   b  of the third frequency conversion part  140  are coupled to each other and connected to the reference signal input part  110 . The drains of the first and second MOS transistors Tr 3   a  and Tr 3   b  are connected to the second output port BB + .  
      The fourth frequency conversion part  150  is connected to the drain of the second MOS transistor Trb of the reference signal input part  110 . The fourth frequency conversion part  150  includes a pair of first and second MOS transistors Tr 4   a  and Tr 4   b  connected in a differential amplifier form. The local oscillator signal LO 270  is input to the gate of the first MOS transistor Tr 4   a  of the fourth frequency conversion part  150  and the local oscillator signal LO 90  having a differential phase difference from the local oscillator signal LO 270  is input to the gate of the second MOS transistor Tr 4   b . The sources of the first and second MOS transistors Tr 4   a  and Tr 4   b  of the fourth frequency conversion part  150  are coupled to each other and connected to the reference signal input part  110 . The drains of the first and second MOS transistors Tr 4   a  and Tr 4   b  are connected to the first output port BB − . That is, the drain of the first frequency conversion part  120  is connected to the drain of the fourth frequency conversion part  140 , and the drain of the second frequency conversion part  130  is connected to the drain of the third frequency conversion part  140 .  
      Each of the first, second, third and fourth frequency conversion parts  120 ,  130 ,  140  and  150  outputs the sum of the reference signal RF and local oscillator signal LO or the difference between the two signals through the first or second output port BB −  or BB + . Here, LO 90 , LO 270 , LO 0 , LO 180  represent phases of the local oscillator signal.  
      The operation of the frequency converter having the aforementioned configuration is explained below.  
      The frequency converter of the present invention can restrain second order intermodulation distortion components through the reference signal input part  110 .  
      The reference signal input part  110  is parallel with the first and second frequency conversion parts  120  and  130  to which differential phases LO 0  and LO 180  are input. The reference signal input part  110  is parallel with the third and fourth frequency conversion parts  140  and  150  to which differential phases LO 90  and LO 270  are input. When the drains of all the MOS transistors of the first, second, third and fourth frequency conversion parts  120 ,  130 ,  140  and  150  are connected, each frequency conversion part has odd symmetrical characteristic. Accordingly, each frequency conversion part restrains the intermodulation distortion products with even-order harmonics of the local oscillator signal LO including self-mixing of the local oscillator signal LO. When the drains of the MOS transistors of the first and fourth frequency conversion parts  120  and  150 , which have an orthogonal phase difference for the local oscillator signal LO, are connected with each other and the drains of the MOS transistors of the second and third frequency conversion parts  130  and  140 , which have an orthogonal phase difference for the local oscillator signal LO, are connected with each other, the output ports BB+ and BB− of the frequency conversion parts  120 ,  130 ,  140  and  150  can suppress the spurious response with a quadruple-order local oscillator signal component and a signal having a first-order harmonic of the reference signal RF and a first-order harmonic of the local oscillator signal LO.  
      When the reference signal RF is sin ω RF t and the local oscillator signal LO is sin ω LO t, the output voltage V BB (t) is represented as follows. 
 
 V   BB ( t )=(3α 5 +5α 5 )sin(ω RF −2ω LO ) t− (3α 3 +5α 5 )sin(ω RF +2ω LO ) t   [Equation 3]
 
       FIG. 5  shows the output spectrum of the even harmonic mixer according to the present invention. Referring to Equation 3 and  FIG. 5 , all of the spurious signals except for a desired signal and a mirror signal are not appear in the output of the even harmonic mixer of the invention.  
      When two closely spaced input tones ω a  and ω b  are input to the RF ports of the reference signal input part, the output voltage V BB (t) is represented as follows. 
 
 V   BB ( t )=α 3  {sin(2ω α +ω LO ) t+ sin(2ω α +ω LO ) t+ sin(2ω b −ω LO ) t+ sin(2ω b +ω LO ) t}   [Equation 4]
 
      It can be known from Equation 4 that the even harmonic mixer of the invention does not all of the intermodulation distortion products include third order intermodulation distortion product and second order intermodulation distortion product. Accordingly, the even harmonic mixer having excellent linearity can be obtained.  
      While the frequency conversion parts and reference signal input part include the MOS transistors in the above-described embodiment, bipolar transistors can replace the MOS transistors as shown in  FIG. 6 .  
      As described above, the even harmonic mixer according to the present invention can remove a DC offset due to self-mixing of a local oscillator signal and second order intermodulation distortion components. Furthermore, while the conventional even harmonic mixer uses only the differential phase of the local oscillator signal, the even harmonic mixer of the invention uses the orthogonal phase difference of the local oscillator signal in addition to the differential phase difference. Thus, the even harmonic mixer of the invention can remove unnecessary output spurious response and intermodulation distortion products so as to obtain excellent output spectrum characteristic and linearity.  
      While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it will be understood by those of ordinary skill in the art that various changes in form and details may be made therein without departing from the spirit and scope of the present invention as defined by the following claims.