Patent Publication Number: US-2023155515-A1

Title: A dc-dc converter assembly

Description:
The present invention relates to a DC-DC converter assembly which comprises a DC-DC power converter configured to convert a DC input voltage, supplied by a DC input voltage source, into a DC output voltage in accordance with a modulated control signal. A converter load electrically connected in series with the DC-DC power converter and the latter comprises a configurable switch network configured to switch the DC-DC power converter between first and second operational modes. 
     BACKGROUND OF THE INVENTION 
     Active and passive components of existing DC-DC power converters are subjected to large voltage and current stresses and large heat dissipation caused by flow of power through the power converter and into the converter load. This reduces reliability and lifetime of DC-DC power converters, in particular high power converters, and/or requires costly active and passive components that can withstand the high currents and/or voltages. Hence, it is desirable to reduce the current stress and/or voltage stress of active and passive components of DC-DC converters of DC-DC converter assemblies for a given or nominal load power. Another disadvantage of existing DC-DC converter assemblies that uses series connection of the converter load and DC-DC power converter is a lacking support of converter load voltages that are both smaller and larger than DC input voltage of the DC-DC power converter. The latter feature requires that the DC-DC power converter is configured or designed to generate both positive and negative the DC output voltages, and also zero for that matter, as discussed in additional detail below. 
     SUMMARY OF THE INVENTION 
     A first aspect of the invention relates to a DC-DC converter assembly which comprises a DC-DC power converter configured to convert a DC input voltage supplied by a DC input voltage source into a DC output voltage in accordance with at least first and second modulated control signals,
         a converter load electrically connected in series with the DC-DC power converter such that the DC input voltage source supplies power directly to the converter load without passing through the DC-DC power converter; said DC-DC power converter comprising:   a control circuit configured to adjust the DC output voltage or current in accordance with a target DC voltage or a DC target current, respectively,   a configurable switch network configured to switch the DC-DC power converter between:   a first operational mode for generating a DC output voltage of a first polarity using a first current charge path and first current discharge path of the configurable switch network to provide a converter load voltage smaller than the DC input voltage, and   a second operational mode for generating a DC output voltage of a second polarity, opposite to the first polarity, using a second current charge path and second current discharge path of the configurable switch network to provide a converter load voltage that is larger than the DC input voltage.       

     By connecting the converter load of the converter assembly in series with the DC-DC power converter, the DC input voltage source may supply a majority of the power delivered into the converter load, for example more than 50%, or more than 66%, or even the substantially entire load power, directly to the converter load. This feature serves to markedly reduce the amount of power that is converted or processed by, i.e. flowing through, the DC-DC power converter for a given or power delivery to the converter load. The ratio between the power supplied directly to the converter load by the DC input voltage source and the power flowing through the DC-DC converter depends on a difference between the desired converter load voltage and the DC input voltage where a small voltage difference leads to large portion of the converter load is delivered directly by the DC input voltage source as discussed below with reference to under the appended drawings. 
     For mains connected applications, the DC input voltage may lie between 320 V and 800 V—for example higher than 565 V. The DC output voltage may be smaller than one-fifth or one-tenth of the DC input voltage for example about 48 V for rechargeable battery pack loads. The load power may be larger than 10 kW or larger than 50 kW. 
     The control circuit may comprise, or form a component or function of, a voltage or current output regulation loop, e.g. based on feedback, that is configured to adjust the DC output voltage, Vout, or DC output current in accordance with the respective target DC voltage or current, the DC input voltage and DC output voltage. The output regulation loop ensures that the DC output voltage or current is dynamically adjusted to maintain a desired or target converter load voltage or current. The output regulation loop ensures that a voltage drop across the converter load is relatively constant and well defined. The control circuit may apply various known control mechanisms to the voltage or current output regulation loop such as pulse width modulation (PWM), phase shift modulation (PSM) or frequency modulation (FM) of the modulated control signal applied to the configurable switch network. 
     In some embodiments, the control circuit is configured to switch the configurable switch network between the first operational mode and second operational mode depending on anyone of the polarity of the DC output voltage, the target DC voltage, the target DC current, the DC input voltage and DC output voltage. 
     In some embodiments, the configurable switch network comprises a plurality of interconnected individually controllable semiconductor switches configured to: 
     during the first operational mode: 
     selectively charge an inductor from the DC output voltage through a first controllable semiconductor switch in accordance with the first modulated control signal (ϕ 1 ) and discharge the inductor into the input of the DC-DC power converter through a second controllable semiconductor switch in accordance with the second, and complimentary, modulated control signal (ϕ 2 ); 
     placing a third controllable semiconductor switch constantly in a non-conducting state; and 
     during the second operational mode: 
     placing the first and second controllable semiconductor switches constantly in a conducting state and non-conducting state, respectively, 
     charge the inductor from the DC input voltage through the third controllable semi-conductor switch in accordance with the one of the first or second complimentary modulated control signals (ϕ 1 , ϕ 2 ) and discharge the inductor into the first or second polarity output of the DC-DC power converter in accordance with the other one of first and second complimentary modulated control signals. 
     The control circuit may comprise a modulator, e.g. pulse width modulator, configured to generate at least the first or second complimentary modulated control signals (ϕ 1 , ϕ 2 ) at respective outputs of first and second comparators of the modulator. The modulator preferably comprises: 
     a carrier signal generator configured to generate first and second mutually offset carrier signals at a switching frequency of the DC-DC power converter. The first comparator may have a first input connected to the first carrier signal and second input connected to a dynamic reference signal and the second comparator may have a first input connected to the second carrier signal and a second input connected to the dynamic reference signal. The control circuit is preferably configured to switch between first and second operational modes by adjusting a voltage or level of the dynamic reference signal as explained in additional detail below with reference to the appended drawings. 
     The characteristics of the dynamic reference signal may be utilized to set or determine a maximum duty cycle value, e.g. below 90% or below 95%, of the first and/or second complimentary modulated control signals and likewise may be utilized set a minimum duty cycle value, e.g. larger than 5% or 10%, of the first and/or second complimentary modulated control signals as discussed in additional detail below with reference to the appended drawings. 
     The control circuit may for example be configured to adjust the voltage or level of the dynamic reference signal at a predetermined control frequency f z , within a predetermined intermediate output voltage region around 0 V of the DC-DC power converter. The control frequency f z , may be at least 3 or 5 times smaller, e.g. between 5 and 10 or times smaller, than the switching frequency of the DC-DC power converter. Hence, if the switching frequency of the DC-DC power converter is 100 kHz, the control frequency f z  is preferably smaller than 33.3 kHz. 
     In some embodiments, the DC-DC power converter comprises: 
     at least one capacitor connected between the positive input and positive output of the DC-DC power converter; or 
     an input capacitor connected between the positive input and negative input of the DC-DC power converter and an output capacitor connected between the positive output and negative output of the DC-DC power converter. 
     In some embodiments, the DC-DC power converter comprises: 
     at least one capacitor connected between the positive input and positive output of the DC-DC power converter; or 
     an input capacitor connected between the positive input and negative input of the DC-DC power converter and an output capacitor connected between the positive output and negative output of the DC-DC power converter. 
     In some embodiments, the configurable switch network is electrically connected between an input and an output of the DC-DC power converter. 
     In some embodiments, the configurable switch network further comprises: 
     a first passive diode connected in series with the inductor and the first controllable semiconductor switch between the positive and negative outputs of the DC-DC power converter to provide, during the first and second operational modes, a first charge path for charging the inductor in accordance with the first modulated control signal (ϕ 1 ); 
     a second passive diode coupled in series with the inductor and the second controllable semiconductor switch to the positive input the DC-DC power converter to provide, during at least the first operational mode, the first discharge path for discharging the inductor in accordance with the second modulated control signal (ϕ 2 ). 
     In some embodiments, the configurable switch network further comprises: 
     a first active diode, for example comprising a fourth controllable semiconductor switch, connected in series with the inductor and the first controllable semiconductor switch between the positive and negative outputs of the DC-DC power converter to provide, during the first and second operational modes, a first charge path for charging the inductor in accordance with first modulated control signal (ϕ 1 ); 
     a second active diode, for example comprising a fifth controllable semiconductor switch, coupled in series with the inductor and the second controllable semiconductor switch to the positive input the DC-DC power converter to provide, during at least the first operational mode, a first discharge path for discharging the inductor during the second phase (ϕ 2 ) of the modulated control signal. 
     In some embodiments, the DC-DC power converter further comprises: 
     a resonant DC-DC converter stage coupled in series with the positive input the DC-DC power converter and configured to step-up the DC input voltage with a predetermined boost factor. 
     In some embodiments, the resonant DC-DC converter stage comprises: 
     a first full-bridge or half-bridge rectifier coupled between the DC input voltage of the DC-DC power converter and a primary side winding of a transformer; 
     a second full-bridge or half-bridge rectifier coupled between a secondary side winding of the transformer and the input voltage of the configurable switch network. 
     In some embodiments, at least one of the converter load and the DC input voltage source comprises an inverter, e.g. AC-DC converter, or a battery pack with a plurality of rechargeable battery cells. 
     In some embodiments, the converter load comprises and the DC input voltage source comprises an inverter, e.g. AC-DC converter, connectable to a single phase mains grid or a three phase mains grid. 
     In some embodiments, the DC-DC power converter is configured for bidirectional operation to additionally transfer power from the converter load directly to the DC input voltage source without passing through the power DC-DC converter. 
     A second aspect of the invention relates to a method of supplying power to a converter load of a DC-DC converter assembly using a DC-DC power converter, comprising:
         connecting a DC input voltage source to an input of the DC-DC power converter to provide a DC input voltage thereto,   adjusting a DC output voltage or current at the output of the DC-DC power converter by a control circuit in accordance with the DC input voltage, the DC output voltage and a target DC voltage or target DC current, respectively;       

     selectively switching a configurable switch network of the DC-DC power converter between: 
     a first operational mode for generating a first polarity DC output voltage to provide a converter load voltage smaller than the DC input voltage; and 
     a second operational mode for generating a second polarity, opposite to the first polarity, DC output voltage to provide a converter load voltage larger than the DC input voltage. 
     In some embodiments, the method further comprises: 
     during the first operational mode: charge an inductor from the DC output voltage through a first controllable semiconductor switch in accordance with a first modulated control signal (ϕ 1 ) and discharge the inductor into the input of the DC-DC power converter in accordance with a second, and complimentary, modulated control signal (ϕ 2 ); 
     switching a third controllable semiconductor switch constantly to a non-conducting state; and 
     during the second operational mode: maintain the first and second controllable semiconductor switches constantly in a conducting state and non-conducting state, respectively, 
     charge the inductor from the DC input voltage through the third controllable semiconductor switch in accordance with the first modulated control signal (ϕ 1 ) and discharge the inductor, through the negative output, into the first or second polarity output of the DC-DC power converter in accordance with the second modulated control signal (ϕ 2 ). 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Preferred embodiments of the invention are described in more detail in the following in connection with the appended drawings, in which: 
         FIG.  1    is a block diagram of an exemplary DC-DC converter assembly in accordance with various embodiments of the invention, 
         FIG.  1 A  schematically illustrates different exemplary embodiments of a DC-DC converter assembly as disclosed herein, 
         FIG.  2    shows a schematic circuit diagram of a first embodiment of a DC-DC power converter of the DC-DC converter assembly, 
         FIG.  3    shows a schematic circuit diagram of the first embodiment of the DC-DC power converter arranged in a first operational state, 
         FIG.  4    shows a schematic circuit diagram of the first embodiment of the DC-DC power converter arranged in a second operational state, 
         FIG.  5    shows a schematic circuit diagram of a second embodiment of a DC-DC power converter of the DC-DC converter assembly, 
         FIG.  6    shows two plots of various voltages, such the converter load voltage, of the DC-DC converter assembly over time, 
         FIG.  7    shows a schematic circuit diagram of a third embodiment of the DC-DC power converter of the DC-DC converter assembly, 
         FIG.  8    shows two plots over time of converter load current, and various voltages, of the DC-DC converter assembly according to the third embodiment thereof, 
         FIG.  9    shows a schematic circuit diagram of a fourth embodiment of the DC-DC power converter of the DC-DC converter assembly, 
         FIG.  10    shows plots over time of converter load current, and various converter voltages, of the DC-DC converter assembly according to the fourth embodiment thereof; and 
         FIG.  11    shows plots of respective control signals over time of a plurality of controllable semiconductor switches of the configurable switch network of the DC-DC power converter according to the fourth embodiment thereof, 
         FIG.  11 A  shows an exemplary embodiment of a modulator circuit of a control circuit of the DC-DC converter assembly, 
         FIG.  11 B  shows carrier signals and first and second sets of modulated control signals generated by the modulator circuit for application to switches of the configurable switch network, 
         FIGS.  12 A,  12 B  show current charge and current discharge paths through the third embodiment of the DC-DC converter assembly in the first operational mode when the load current Iload is negative, 
         FIGS.  13 A,  13 B  show current charge and current discharge paths through the third embodiment of the DC-DC converter assembly in the first operational mode when the load current Iload is positive, 
         FIGS.  14 A,  14 B  show current charge and current discharge paths through the third embodiment of the DC-DC converter assembly in the second operational mode at negative load current Iload; and 
         FIGS.  15 A,  15 B  show current charge and current discharge paths through the third embodiment of the DC-DC converter assembly in the second operational mode at positive load current Iload. 
     
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     The following section describes various exemplary embodiments of the present DC-DC converter assembly with reference to the appended drawings. The skilled person will understand that the accompanying drawings are schematic and simplified for clarity and therefore merely show details which are essential to the understanding of the invention, while other details have been left out. Like reference numerals refer to like elements or components throughout. Like elements or components will therefore not necessarily be described in detail with respect to each figure. The skilled person will further appreciate that certain actions and/or steps may be described or depicted in a particular order of occurrence while those skilled in the art will understand that such specificity with respect to sequence is not actually required. 
       FIG.  1    shows a schematic diagram of an exemplary DC-DC converter assembly  100  in accordance with the invention. The DC-DC converter assembly  100  comprises a DC-DC power converter  101  which converts a certain portion or percentage of a load power supplied to a converter load  110  (Load/Source), while a DC input voltage source or current source  120  (Source/Load) supplies the residual portion of the load power directly to the converter load  110  without passing through the DC-DC power converter  101 . The direct supply of load power to the converter load  110  is achieved because the converter load  110  is electrically connected between a positive input  103  and positive output  108  of the DC-DC power converter  101 —for example via an electrical wire or conductor  112 . This load connection arrangement connects the converter load  110  in series with the DC-DC power converter  101  instead of the traditional parallel connection of the converter load to the positive and negative outputs  108 ,  107 . In some embodiments of the DC-DC converter assembly  100 , the load power delivered directly to the converter load  110  by the DC input voltage source  120  may be markedly larger than the load power delivered by DC-DC power converter  101 —for example at least 2, 3, 5 or 10 times larger depending on design details, requirements to the converter load voltage and the DC input voltage supplied by DC input voltage source  120  and certain performance requirements of the DC-DC converter assembly  100 . 
     The reduction of power delivery by the DC-DC power converter  101  may leads to considerable reduction in size and costs of the DC-DC power converter  101  at a specified load power. The reduced power delivery requirements to the DC-DC power converter  101  entail further advantages such as increased reliability because voltage stress and heat dissipation of active and passive components of the DC-DC core  102  are reduced and component costs are reduced. The overall energy/power efficiency of the DC-DC converter assembly  100  is also increased because the DC-DC power converter  101  converts less power and accordingly reduces power losses within the converter core  102 . The DC input voltage source  120  is preferably connected between a positive input  103  and negative input  104  of the DC-DC power converter  101 . The negative input  104  may for example be connected to a ground potential of the DC converter assembly  100  and a negative output  107  also connected to the ground potential. 
     The DC-DC power converter  101  additionally comprises a control circuit  113  configured to adjust the DC output voltage, Vout, at the output terminal  122  in accordance with a target DC voltage, Vref, or equivalent adjusting a DC output current, or output power, flowing through the output terminal  122  in accordance with a target DC current. The control circuit  113  preferably forms a component or function of a voltage or current feedback regulation loop that is configured to adjust the DC output voltage, Vout, or DC output current in accordance with the respective target DC voltage or current  111   c,  the current DC input voltage Vin and current DC output voltage Vout. The control circuit  113  generates a set of control signals  111  which controls the respective state switchings of a plurality of controllable semiconductor switches of the DC/DC core and sets a switching frequency of the DC-DC power converter  101 , for example a frequency between 10 kHz to 1 MHz as discussed in additional detail below. The skilled person will understand that control circuit may use various types of modulation of the modulated control signal or signals  111  such as pulse width modulation (PWM), PSM, PDM or FM. 
     The target DC voltage, Vref, preferably represents a desired converter load voltage, Vload, and the control circuit  113  is configured to monitor or determine the DC output voltage, Vout, and the DC input voltage, Vin, to make appropriate adjustment of the converter load voltage, Vload, because the latter voltage is the difference between Vin and Vout due to the series connection of the converter load  110  and the DC-DC power converter  101  in the converter assembly  100 . The control circuit  113  is preferably configured to seamlessly and dynamically, under normal operation of the assembly  100 , switch between the first operational mode and the second operational mode of the power converter  101  depending on a target converter load voltage Vload and the current DC input voltage Vin. 
     The skilled person will appreciate that some embodiments of the DC-DC power converter  101  may be unidirectional where power only can be transferred from the source  120  to the converter load  110 . Such unidirectional DC-DC converters may comprise a passive rectification circuit. Alternative embodiments of the DC-DC converter  101  may be bidirectional enabling power transfer from the source  120  to the load  110  and vice versa depending on a suitable control mechanism applied to an active rectification circuit on the secondary side as described in additional detail below. In the latter embodiments, the skilled person will understand that the role of the DC input voltage source  120  and the converter load  110  in practice may be interchanged when the DC-DC power converter  101  operates in reverse mode where the DC input voltage source  120 , as indicated by the “Load” designation receives power from the converter load  110  which therefore also is indicated as “Source”. This feature is particularly advantageous for example where the converter load, or the DC input voltage source  120 , comprises a battery pack  120   c,  e.g. including a plurality of rechargeable battery cells. In the latter embodiment, energy stored in the battery cells may be supplied to the converter load, e.g. an AC motor of an EV, and energy generated by the AC motor in reverse operation e.g. a regenerative mode, may be used to charge the battery cells. The DC input voltage source may comprise a two-phase or three-phase grid-connected inverter  120   a  and/or the converter load  110  may comprise an energy storage unit such as a rechargeable battery stack or package comprising a plurality of series connected rechargeable battery cells or a fuel cell etc. The converter load  110  may alternatively comprise a regenerative fuel cell (RFC) or a grid-connected inverter such that the grid acts as a converter load and the energy storage unit may deliver power/energy to the grid for example for grid stabilization purposes or deliver power/energy to AC loads such as dishwashers or washing machines. 
     The absolute value of the DC output voltage Vout, as set by the control circuit  113 , may be significant smaller than the absolute value of the DC input voltage, Vdc, supplied by the DC input voltage source  120  at the positive and negative inputs  103 ,  104 , of the DC-DC power converter  101 . This feature ensures that the majority of the load power is supplied by the DC input source  120  as illustrated by the quantitative example below. 
     One exemplary embodiment of the DC-DC converter assembly  100  may be designed or constructed using the following constraints and target performance: 
     
       
         
           
               
               
             
               
                 TABLE 1 
               
               
                   
               
               
                 Vdc &gt; Vload 
                 Vdc &lt; Vload 
               
               
                   
               
             
            
               
                 Vdc = 50 V 
                 Vdc = 46 V 
               
               
                 Vload = 48 V 
                 Vload = 48 V 
               
               
                 Pload: = 1 kW 
                 Pload: = 1 kW 
               
               
                 Iload = 1 kW/48 V = 20.83 A 
                 Iload = 1 kW/48 V = 20.83 A 
               
               
                 Vout = Vdc − Vioad = 50 V − 
                 Vout = Vdc − Vioad = 46 V − 
               
               
                 48 V = 2 V 
                 48 V = −2 V 
               
               
                 Iout = Iload = 20.83 A 
                 Iout = Iload = 20.83A 
               
               
                 Pconverter = Vout * Iout = 2 V * 
                 Pconverter = Vout * Iout = −2 V * 
               
               
                 20.83 A = 41.67 W 
                 20.83 A = 41.67 W 
               
               
                 Fsw = 100 kHz. 
               
               
                 Inductance of Inductor L = 1 microH 
               
               
                   
               
            
           
         
       
     
     Consequently, in the above design example of the DC-DC converter assembly  100  the DC input voltage source Vdc supplies about 958 W directly to the converter load  110  while the residual 42 W of the total 1 kW load power is supplied by the DC-DC power converter  101 . Hence, demonstrating a marked reduction of power supplied by or through the DC-DC power converter  101  compared to conventional solutions where the converter load is coupled in parallel to the output of DC-DC power converter  101 . 
     The skilled person will understand that the design specification of the above-mentioned exemplary embodiment of the DC-DC converter assembly  100  requires that the DC-DC power converter  101  is configured to, or capable of, generating a positive DC output voltage Vout in order to supply a converter load voltage Vload that is smaller than the DC input voltage, equal to Vdc, of the DC-DC power converter  101 . However, the design specification additionally requires that the DC-DC power converter  101  is configured to, or capable of, generating a negative DC output voltage Vout in order to supply a converter load voltage Vload that is larger than the DC input voltage, equal to Vdc, of the DC-DC power converter  101 . This feature implies that the DC-DC power converter  101  is capable of operating in boost mode as well as buck mode which on one hand improves the flexibility of the converter assembly  100  and on the other hand for certain cases, the rated power of the DC-DC power converter  101  can be halved compared to a corresponding power converter which only functions where the converter load voltage Vload is smaller than the DC input voltage. Several DC-DC power converter embodiments or topologies with this advantageous boost mode and buck mode capability is discussed below in detail. 
       FIG.  1 A  schematically illustrates different exemplary embodiments of a DC-DC converter assembly as disclosed herein. Shown are four different exemplary serial couplings between power converter and load. 
       FIG.  2    shows a simplified circuit diagram of a first embodiment of the DC-DC power converter  101  of the DC-DC converter assembly  100  without details of the control circuit  113  for brevity. The converter load, Rload, and DC input voltage source/generator Vdc associated with the converter assembly  200  are included as well to clarify the interconnections. The DC-DC power converter  201  comprises a configurable switch network electrically connected between the positive input Vin+ and the positive output, Vout+, of the DC-DC power converter  201 . The configurable switch network comprises a plurality of interconnected individually controllable semiconductor switches S 1 , S 2  and S 3  that are switched between conducting and non-conducting states, i.e. between on and off, by respective control signals Control S 1 - 3  connected to respective gate, or similar device control terminals, supplied by the previously discussed control circuit via a bus or set of wires  211  of the power converter  101 . Each of the controllable semiconductor switches S 1 , S 2  and S 3  may comprise a bipolar transistor or a FET such as a MOSFET or an IGBT. The configurable switch network comprises an inductor L 1  with one end connected to a drain or source terminal of S 3  and to the cathode of diode D 1 . The other end of the inductor L 1  is connected to the drain of S 1  and anode of the diode D 2 . S 3  is connected between the cathode of D 1  and the positive DC input of the power converter  101 . The configurable switch network preferably also comprises an output smoothing capacitor C 1  connected from the DC output Vout to a neutral or ground rail  204  of the power converter  101  and an input smoothing capacitor C 2  connected from the DC voltage input Vin to the neutral or ground rail  204 . 
     The control circuit is configured to, via the control signals Control S 1 - 3 , switch the configurable switch network between a first operational mode and a second operational mode depending on the target or desired converter load voltage Vload, the DC input voltage and DC output voltage. The control circuit is configured or designed to select the first operational mode where the converter load voltage Vload is smaller than the DC input voltage which implies Vout is a positive voltage relative to circuit ground. 
       FIG.  3    shows a an equivalent circuit diagram of the DC-DC power converter  201  arranged in the first operational mode where the third controllable semiconductor switch S 3  is switched constantly to its off/non-conducting state, i.e. an open connection. The gate terminals of the first and second controllable semiconductor switches S 1 , S 2  are driven by Control S 1  and S 2 , respectively, which are complementary of the modulated control signal—for example PWM modulated control signals to provide duty-cycle based adjustment of Vout. Thereby, in the first operational mode of the configurable switch network the inductor L 1  is charged from Vout through the first controllable semiconductor switch S 1 , and through the forward biased diode D 1 , in accordance with the first control signal PWM 1  or (ϕ 1 ) of the complementary modulated control signals PWM 1 , PWM 2 . When the second control signal PWM 2 , which is complementary of PWM 1 , is active or logic high switch S 1  is non-conducting while switch S 2  is switched to its on/conducting state such that the current flowing through the inductor L 1  is redirected into the positive input  203  of the DC-DC power converter  101  by passing through the conducting, and therefore low-resistance, state of S 2 . Accordingly, power is transferred from the output  208  to the input  203  of the DC-DC power converter  201  in its first operational state. 
     A more detailed explanation of the use of a first current charge path and current discharge path in the first operational state of the exemplary DC-DC power converter, that may be identical to the third embodiment of the DC-DC power converter  701 , follows here: 
     If the DC input voltage Vdc&gt;Vload, the DC-DC power converter  701  operates in boost mode corresponding to the first operational mode. The configurable switch network comprises five individually controllable semiconductor switches S 1 , S 2 , S 2 ′, S 3  and S 4  in the present embodiment of the power converter  701  while other embodiments may comprise fewer or additional controllable switches. Switches S 3  and S 4  are driven by complementary gate control signals that are preferably supplied by the control circuit such that switch S 4  is always ON or conducting, and switch S 3  is always OFF during, or in, the first operational mode. Switches S 1  and S 2  are driven by the complementary PWM gate control signals while switches S 2  and S 2 ′ are driven by identical modulated gate control signals. The analysis below is based on inductor current continuous conduction mode (CCM) of the DC-DC power converter  701 . When the load current Iload is negative, i.e. discharging the battery load through DC-DC power converter  701  into the DC input voltage source Vs, the control circuit turns on switch S 2  and switch S 2 ′—Thereby, the inductor L 1  is charged via the first current charge path. Thereafter, the control circuit turns off switches S 2  and S 2 ′, which induces an inductor current that is freewheeling through the first discharge current path through switch S 1 . The DC input voltage Vin supplied by DC input voltage source Vs is bucked down to the DC output voltage Vout. The first charge current path and discharge current paths are schematically illustrated on  FIGS.  12 A,  12 B  even though only the current paths inside the DC-DC power converter  701  are illustrated for simplicity. 
       FIGS.  12 A,  12 B  show current charge and discharge paths, respectively, when the load current Iload is negative in the first operational mode. 
       FIG.  12 A  illustrates how switches S 2 , S 2 ′, S 4  are ON while switches S 1  and S 3  are OFF.  FIG.  12 B  illustrates how switches S 1  and S 4  are ON while switches S 2 , S 2 ′ and S 3  are OFF. 
     At positive load currents Iload in the first operational mode of the power converter  701  and assembly, i.e. charging the exemplary rechargeable battery pack load through DC-DC power converter  701  from the DC input voltage source Vs, the control circuit turns on switch S 1  and the inductor L 1  is charged through a third current charge path. The control circuit thereafter turns off switch S 1  and turns on switches S 2  and S 2 ′ which serve to discharge the inductor current through a third current discharge path which includes the conducting, or ON-state, switches S 2  and S 2 ′. This discharge action in turn boosts the voltage up from Vout to Vs. These current charge and discharge paths are shown on  FIGS.  13 A,  13 B  even though only the current paths inside the DC-DC power converter  701  are illustrated for simplicity. 
       FIGS.  13 A,  13 B  illustrate the charge and discharge current paths when the load current is positive in the first operational mode.  FIG.  13 A  shows that S 1  is ON and  FIG.  13 B  shows that S 2  and S 2 ′ are ON and S 4  is ON while S 1  is OFF. 
     In response to the DC input voltage Vs, corresponding to Vin, is smaller than the converter load voltage, Vload, the control circuit switches the DC-DC power converter  701  into the second operational mode which may comprise a buck-boost mode. Switches S 3  and S 4  are driven by the complementary modulated, e.g. PWM, control or gate drive signals. Switches S 1  and S 2  are driven by complementary modulated control or gate drive signals. Switches S 2  and S 2 ′ are preferably driven by the same gate driving signals. Switch S 1  is preferably always ON during the second operational mode of the power converter while switches S 2  and S 2 ′ are preferably always OFF or non-conducting during the second operational mode. The analysis below is based on a preferred inductor current continuous conduction mode (CCM) of the power converter  701 . 
     When the load current !load is negative, i.e. discharging the rechargeable battery pack or cell based converter load through DC-DC converter  701  to the input voltage Vs, the control circuit is configured to turn ON switch S 4  such that inductor L 1  is charged through a second current charge path. The control circuit subsequently turns off switch S 4  and turns on switch S 3  that leads to a discharge of the inductor current through a second discharge path as illustrated on  FIGS.  14 A,  14 B . At the same time, the inductor discharge current charges capacitor C 2  from top to bottom. The current paths are shown in  FIGS.  14 A,  14 B  albeit only the charge and discharge current paths inside the DC-DC converter  701  are illustrated for simplicity. 
       FIGS.  14 A,  14 B  show charge current and discharge current paths in the second operational mode when the load current Iload is negative.  FIG.  14 A  shows that switch S 4  is on while  FIG.  14 B  shows that switch S 3  is on or conducting. 
     When the load current is positive, i.e. charging the battery through the DC-DC converter from Vs, the control circuit may turn on S 3  to charge L 1  in response; the control circuit may thereafter turn off S 3  and turn on S 4  such that inductor current in L 1  is discharged through S 4  and charges capacitor C 1  (from bottom to top). The current paths are shown in  FIG.  14 A , B (only the current paths in the DC-Dc converter are illustrated for brevity. 
       FIGS.  15 A,  15 B  illustrate charge current and discharge current paths in the second operational mode when the load current Iload is positive.  FIG.  15 A  shows that switch S 3  is on and  FIG.  15 B  shows that switch S 4  is on. 
     Going back to the DC-DC power converter of  FIG.  3    and  FIG.  4   , the control circuit is configured or designed to select the second operational mode in response to the converter load voltage Vload exceeds the DC input voltage. The latter condition implies that Vout is a negative voltage relative to circuit ground  204 . 
       FIG.  4    shows an equivalent circuit diagram of the DC-DC power converter  201  arranged in the second operational mode where the first controllable semiconductor switch S 1  is switched constantly to its on/conducting state, i.e. effectively acting as a short. The second controllable semiconductor switch S 2  is switched constantly to its off/non-conducting state, i.e. effectively acting as an open circuit. The gate terminal of the third controllable semiconductor switch S 3  is driven by Control S 3  that is one of the complementary modulated control signals PWM 1 , PWM 2  of the previously discussed modulated control signal. Thereby, in the second operational mode of the configurable switch network, when the switch S 3  is on the inductor L 1  is charged through a first current charge path running from the DC input voltage Vin through the small on-resistance of S 3 , and through a small on-resistance of the conducting switch S 1 . When the switch S 3  is switched to its off/non-conducting state the current flowing through the inductor L 1  is redirected, or discharged through a first current discharging path running through the ground connection  204  into the output capacitor C 1  and into the positive output  208  of the DC-DC power converter as illustrated by current flow path IL 1  so as to decreased the DC output voltage Vout. Accordingly, power is transferred from the input  203  to the output  208  of the DC-DC power converter  201  in its second operational state and the power converter is operated in boost-buck mode. 
       FIG.  5    shows a simplified circuit diagram of a second embodiment of the DC-DC power converter  101  of the DC-DC converter assembly  100  without details of the control circuit  113  for brevity. The converter load, Rload, and DC input voltage source/generator Vdc associated with the converter assembly  300  are included as well to clarify the interconnections. The functionality and topology of the DC-DC power converter  301  is largely identical to the previously discussed DC-DC power converter  201  except for the number and coupling of the input and output smoothing capacitors C 1 , C 2 . The latter smoothing capacitors are replaced by a single so-called flying capacitor C 3  which is interconnected between the input Vin and output Vout of the DC-DC power converter  301 . 
     The upper plot  602  of  FIG.  6    shows a simulation of voltages and currents of the above-discussed first embodiment of the DC-DC converter assembly  200 ,  300 ,  700  operating in the first operational mode in response to the converter load voltage Vload is smaller than the DC input voltage Vin of the power converter  201 . 
     The skilled person will understand that actual component values depend on target performance of the DC-DC power converter assembly, in particular current ripple/voltage ripple specifications. In certain useful embodiments of the DC-DC converter assembly  200 ,  300 ,  700  the switching frequency fsw may be between 50 kHz and 200 kHz, for example using C 1 =C 2 =47 μF, L 1 =100 μH. 
     As illustrated, the converter load voltage Vload is set to about 38 V and the DC input voltage Vin, as supplied by the DC input voltage source Vdc, is about 48 V which mean that the DC output voltage Vout of the converter is about 10 V in steady state operation after initial settling. In some embodiments (with inductor current being about 12.8A, Vload being about 39.2V, and Vout being about 8.9V, the power delivery ratio between DC input source and converter output is about 5 to 1. 
     The lowermost plot  604  of  FIG.  6    shows a simulation of voltages and currents of the above-discussed first embodiment of the DC-DC converter assembly  200  operating in the second operational mode where the converter load voltage Vload is larger than the DC input voltage Vin of the power converter  201 . As illustrated, the converter load voltage Vload is set to about 58 V and the DC input voltage Vin, as supplied by the DC input voltage source Vdc, is about 48 V which means that the DC output voltage Vout of the converter is about −10 V in steady state operation after initial settling. In some embodiments (with inductor current being about 66.8 A, Vload being about 58.2V, and Vout being about −10.2V), the power delivery ratio between DC input source and converter output is about 6 to 1. 
       FIG.  7    shows a simplified circuit diagram of a third embodiment of the DC-DC power converter  101  of the DC-DC converter assembly  100  without details of the control circuit  113  for brevity. The converter load, Rload, and DC input voltage source/generator Vdc associated with the converter assembly  700  are included as well to clarify the interconnections. The converter load, Rload, may comprise a rechargeable battery bank or stack, Vbat, or similar energy storage element, with a certain internal resistance R_bat as schematically illustrated by the diagram. In numerous important applications, it is advantageous if the DC-DC power converter  701  supports both charging and discharging of the rechargeable battery bank from the DC input voltage source/generator Vdc as illustrated by bi-directional load/battery current Iload and depending on the charging state of the battery bank the converter load voltage by be larger or smaller than the DC input voltage Vin. Therefore, the DC-DC power converter  701  supports full bidirectional operation. The functionality and topology of the DC-DC power converter  701  is largely identical to the previously discussed DC-DC power converter  201  except for a replacement of the diode D 1  with a fourth controllable semiconductor switch S 4  in the configurable switch network and replacement of the diode D 2  with a fifth controllable semiconductor switch S 2 ′. The control circuit is modified accordingly to provide respective control signals, Control S 1 -S 4 , to gate terminals of the switches S 1 -S 4 . 
     When the DC-DC power converter  701  is arranged in, or switched to, its first operational mode, the gate terminals of the first and second controllable semiconductor switches S 1 , S 2  are driven by Control S 1  and S 2 , respectively, which preferably are complementary duty cycle modulated control signal. Furthermore, in the first operational mode, switch S 3  preferably resides constantly in its non-conducting state, i.e. OFF, and switch S 4  resides constantly in its conducting state, i.e. ON. Switches S 2  and S 2 ′ may be driven by identical modulated control signal and identical switching patterns, i.e. S 2  on=S 2 ′ on, S 2  off=S 2 ′ off. Alternatively, a bit more complex switching may be utilized if S 2  is constant ON in the first operational mode. S 3  is constantly non-conducting and S 4  is preferably constantly conducting to provide a charge and/or discharge path for L 1 . The switches S 2  and S 2 ′ may be connected with common drain or alternatively connected with common source. The latter connection may simplify the control circuit of the switches S 2  and S 2 ′ as they could share a single isolated gate driver power supply. 
     When the DC-DC power converter  701  operates in its second operational mode, the gate terminals of the third and fourth switches S 3 , S 4  are driven by Control S 3  and S 4 , respectively, which may be complementary phases (ϕ 1 , ϕ 2 ) of the previously discussed modulated control signal, while switch S 2  resides constantly in its non-conducting state and switch S 1  resides constantly in its conducting state. Switch S 2 ′ should also reside constantly in its non-conducting state because switch S 2  resides constantly in its non-conducting state. 
       FIG.  11 A  shows an exemplary embodiment of a modulator  1113  of the control circuit  113  which may be utilized in any of the previously disclosed DC-DC power converters  201 ,  301 ,  701 . The modulator  1113  is configured to generate the respective modulated control signals, Control S 1 , S 2 , S 2 ′ S 3 , S 4 , to the respective gate terminals of the switches S 1 , S 2 , S 2 ′, S 3  and S 4  in a manner which provides seamless switching between the first and second operational modes of the DC-DC power converter as needed in response to requirements for positive and negative DC output voltages Vout. The seamless switching between the first and second operational modes is achieved through the use of an intermediate output voltage region, as defined below, which comprises a modified switching pattern between the first and second operational modes. The intermediate output voltage region ensures that the DC-DC power converters can dynamically switch between positive and negative load/output currents without generating undesired spikes or noise in Vout or the load current. The intermediate output voltage region is defined as a pre-set output voltage range between small positive and small negative values of Vout, e.g. between +1 V and −1V or similar predetermined upper and lower level thresholds that are substantially smaller than a nominal DC output voltage of the power converter assembly. 
     The control circuit  113  preferably comprises first and second comparators  1120 ,  1122 . The first comparator  1120  generates a first set of pulse width modulated (PWM) control signals, Control S 1 , S 2 , as complimentary signals via an inverter  1123 . The second comparator  1122  generates a second set of pulse width modulated control signals, Control S 3 , S 4 , as complimentary signals via a second inverter  1124 . A first carrier signal, denoted carrier signal # 1 , which may have a frequency corresponding to the previously discussed switching frequency fsw of the DC-DC power converter, is applied to one input of the first comparator  1120 . A dynamic reference signal  1130 , as illustrated on  FIG.  11 B , is applied to the second, or other, input of the first comparator  1120 . A second carrier signal, denoted carrier signal # 2 , is applied to a first input of the second comparator  1122  while the dynamic reference signal  1130  is applied to the second, or other, input of the second comparator  1122 . The first and second carrier signals preferably have identical frequency, peak-peak voltage or current amplitudes and phases, but are mutually DC offset e.g. with a predetermined offset voltage. The predetermined offset voltage preferably corresponds to a peak-peak voltage of the first or second carrier signal, e.g. schematically illustrated as “1” on  FIG.  11 B  but may in a practical circuit implementation correspond to a voltage between 2 and 20 Volts. In this manner, a peak voltage of the first carrier signal is substantially equal to a minimum voltage of the second carrier signal as schematically illustrated by the respective waveforms of the first and second carrier signals on  FIG.  11 B . 
     The respective waveforms of the first and second carrier signals and the dynamic reference signal  1130  of the control circuit  113  within the intermediate output voltage region is schematically illustrated on  FIG.  11 B  by the coloured region under Vout=0. The respective waveforms of the first and second carrier signals and the dynamic reference signal  1130  within a normal output voltage region with large positive Vout using the first operational mode of the power converter is schematically illustrated on  FIG.  11 B  by the left-most coloured region under Vout&gt;0. Likewise, the respective waveforms of the first and second carrier signals and the dynamic reference signal  1130  within a normal output voltage region with large negative Vout, e.g. larger than the predetermined lower level threshold, using the second operational mode of the power converter is schematically illustrated on  FIG.  11 B  by the right-most coloured region under Vout&lt;0. The skilled person will appreciate that the level of the dynamic reference signal  1130  relative to the respective levels of the first and second carrier signals within the normal output voltage region Vout&gt;0 ensures that merely the first set of pulse width modulated control signals, Control S 1 , S 2 , are actively modulated while the second set of pulse width modulated control signals, Control S 3 , S 4 , are inactive to render the switches S 3 , S 4  in constant non-conducting state and conducting state, respectively, as previously discussed. Conversely, the level of the dynamic reference signal  1130  relative to the respective levels of the first and second carrier signals within the normal output voltage region Vout&lt;0 ensures that merely the second set of pulse width modulated control signals, Control S 3 , S 4 , are actively modulated while the first set of pulse width modulated control signals, Control S 1 , S 2 , are inactive to render the corresponding switches S 1 , S 2  in constant conducting state and non-conducting state, respectively. 
     The dynamic reference signal  1130  within the intermediate output voltage region under Vout≈0 is dynamically switched between the mid-level voltage, “1”, with voltage amplitude steps of Δv 1  and Δv 2  at a control frequency of f z,.  The modulation frequency of f z,  may be at least 3 times smaller, e.g. between 5 and 20 times smaller, than the switching frequency of the DC-DC power converter. The control frequency of f z  is preferably higher than 15 kHz or 20 kHz to avoid audible buzz or noise. The size of Δv 1  is utilized to set or determine a maximum duty cycle, e.g. 90% or 95%, of the pulse width modulated control signal Control S 1 . The setting of the maximum duty cycle is helpful to prevent driving the duty cycle into a range close 100% where practical component limitations and parasitics of the switches S 1 -S 4 , comparators  1120 ,  112  etc. make the actual duty cycle uncertain and hence leads to uncontrolled voltage spikes and anomalies in the pulse width modulated control signals Control S 1 -S 4  and/or uncontrolled voltage spikes and anomalies in the output voltage and/or load current of the DC-DC power converter. In a similar manner, the size of Δv 2  is utilized to set or determine a minimum duty cycle, e.g. 5% or 10%, of the pulse width modulated control signal Control S 3  with corresponding advantages. 
     In this manner, the above-mentioned characteristics of the dynamic reference signal  1130  within the intermediate output voltage region, where Vout is close to zero, ensures that the DC-DC power converter toggles in a well-controlled manner between the first and second operational modes at the control frequency of f z . 
     The upper plot  802  of  FIG.  8    shows a simulation of voltages and currents of the above-discussed third embodiment of the DC-DC converter assembly  700  operating in the first operational mode where the converter load voltage Vload is smaller than the DC input voltage Vin of the power converter  701 . As illustrated, the converter load voltage Vload is set to about 48 V and the DC input voltage Vin, as supplied by the DC input voltage source Vdc, is constantly about 50 V which means that the DC output voltage Vout of the converter is about 2.5 V in steady state operation after initial settling. The plot of the converter load current Iload, marked by legend Ibat due to the battery pack load, illustrates the bidirectional load current supply capability of the power converter  701  operating in the first operational mode where Iload in a gradual and well-controlled manner changes direction e.g. transits from negative to positive over time, i.e. from about −25 A at t=0 to about +25 A at t=0.015 s. This current direction switching capability is controlled by the control circuit (not shown) via appropriate control of the respective control signals, Control S 1 -S 4 , to the gate terminals of the switches S 1 -S 4 . 
     The lowermost plot  804  of  FIG.  8    shows a simulation of voltages and currents of the above-discussed third embodiment of the DC-DC converter assembly  700  operating in the second operational mode where the converter load voltage Vload is larger than the DC input voltage Vin of the power converter  701 . As illustrated, the converter load voltage Vload is set to about 51.5−52 V and the DC input voltage Vin, as supplied by the DC input voltage source Vdc, is constantly about 50 V which means that the DC output voltage Vout of the converter is about minus 1.5 V in steady state operation after initial settling. The plot of the converter load current !load, marked by legend Ibat as in plot  802  illustrates the bidirectional load current supply capability of the power converter  701  operating in the second operational mode where Iload gradually and well-controlled changes direction e.g. transits from negative to positive over time, i.e. from about −26 A at t=0 to about +26 A at t=0.015 s. This current direction switching capability is controlled by the control circuit (not shown) via appropriate control of the respective control signals, Control S 1 -S 4 , to the gate terminals of the switches S 1 -S 4 . 
       FIG.  9    shows a simplified circuit diagram of a fourth embodiment of the DC-DC power converter  101  of the DC-DC converter assembly  100  without details of the control circuit  113  for brevity. The converter load, Rload, and DC input voltage source/generator Vdc associated with the converter assembly  900  are included as well to clarify the interconnections. Compared to the first DC-DC power converter embodiment  201  as discussed above, the present DC-DC power converter  901  comprises an additional resonant DC-DC converter stage or circuit  905  connected in series with the positive input  903  the DC-DC power converter  901  and preferably infront of the configurable switch network which, as in the previously discussed embodiments may comprise a plurality of interconnected individually controllable semi-conductor switches S 1 , S 2  and S 3  and diodes D 1 , D 2  or corresponding active diodes. Hence, the functionality and topology of the configurable switch network may be largely identical to anyone of those of the previously discussed DC-DC power converters. 
     The resonant DC-DC converter stage  905  is configured to step-up the DC input voltage with a predetermined boost or buck factor or DC amplification which relaxes boost or buck factor requirements of the configurable switch network. This allows the modulated control signal to operate with a smaller variation of the modulation index relaxing accuracy requirements of the modulated control signals and component stresses of active components of the DC-DC power converter  901 . 
     The resonant DC-DC converter stage  905  is preferably configured to operate in so-called zero voltage switching (ZVS) or zero current switching (ZCS) mode at a resonant frequency of a resonant tank comprising tank inductances Lr 2 , Lm and Lr 1  and tank capacitors Cr 1 , Cr 2 . The ZVS or ZCS mode decreases power dissipation of one or more controllable semiconductor switches S 8 , S 9 , S 10 , S 11 , such as IGBT switches or MOSFET switches, of a full-bridge, or H-bridge, input driver connected to a primary side winding of transformer. The transformer may have a step up ratio, n, between 2 and 100 e.g. between 5 and 25. 
     The resonant DC-DC converter stage  905  comprises a second H-bridge or full-bridge rectifier comprising controllable semiconductor switches S 4 , S 5 , S 6 , S 7  coupled between a secondary side winding of the transformer and an input voltage of the configurable switch network across smoothing capacitor C 2 . The skilled person will understand that the configurable switch network may be driven by modulated control signals  911  that are identical to those of previously discussed modulated control signals  111 ,  211 ,  311 , in particular using the same switching frequency, while the resonant DC-DC converter stage  905  may be operated at the same switching frequency or at a different switching frequency, in particular a switching frequency that maximizes the power or energy efficiency of the resonant DC-DC converter stage  905 . The switching frequency of the resonant DC-DC converter stage  905  may accordingly be set to a frequency at, or close to, the resonant frequency of the resonant tank. 
     The upper plot  1002  of  FIG.  10    shows a simulation of load current Iload in the converter load, Rload, in form of one or more rechargeable batteries or a rechargeable battery pack of the above-discussed fourth embodiment of the DC-DC converter assembly  900  for changing load voltage Vload as illustrated by plot  1004 . As illustrated, using increasing load current leads to increasing load voltage Vload due to the internal resistance of the rechargeable battery or batteries. The increase in load voltage Vload results in a lower output voltage. 
     The lower plot  1004  of  FIG.  10    shows a simulation of voltages and currents of the above-discussed fourth embodiment of the DC-DC converter assembly  900  seamlessly and dynamically switching between the first operational mode, where the converter load voltage Vload is smaller than the DC input voltage Vin of the power converter  901 , and the second operational mode, where the converter load voltage Vload is larger than the DC input voltage Vin. As illustrated, Vin remains fixed at about 50 V over the plotted time span of about 1 s while the converter load voltage Vload, as defined by the previously discussed Vref input to the control circuit  113  (refer to  FIG.  1   ) increases from about 38 V at t=0 to 60 V peaking at t=0.7 s. As illustrated by plot  1004 , the DC output voltage Vout of the converter  901  changes polarity seamless at about t=0.4 S and varies from about from about +15 V at t=0 to about −10 V at t=0.7 s. 
     The upper plot  1102  of  FIG.  11    shows the respective control signals Control S 1 - 4  applied to the individually controllable semiconductor switches S 1 , S 2 , S 3  and S 4  of the configurable switch network of the previously discussed third embodiment of the DC-DC power converter  701  operating in the second operational mode where the converter load voltage Vload is larger than the DC input voltage leading to a negative Vout voltage. As discussed before, in the second operational mode switch S 1  is constantly arranged in its on/conducting state while switch S 2  is constantly in its off/non-conducting state as indicated by the respective levels of the control signals. The modulated control signal Control S 3  is applied to the gate terminal of switch S 3  and a complementary modulated control signal Control S 4 /D 1  is applied to the gate terminal of the switch S 4 . 
     The lower plot  1104  of  FIG.  11    shows the respective control signals Control S 1 - 4  applied to the individually controllable semiconductor switches S 1 , S 2 , S 3  and S 4  of the configurable switch network of the previously discussed third embodiment of the DC-DC power converter  701  operating in the first operational mode where the converter load voltage Vload is smaller than the DC input voltage leading to a positive Vout voltage. As discussed before, in the first operational mode switches S 1  and S 2  are driven by complementary phases (ϕ 1 , ϕ 2 ) of the modulated control signals Control S 1  and S 2 , respectively. The switch S 3  is constantly arranged in its off/non-conducting state and switch S 4  is constantly arranged in its on/conducting state as indicated by the level of the associated gate control signals S 3  and S 4 /D 1 , respectively. 
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