Patent Publication Number: US-2010124326-A1

Title: Subscriber line interface circuitry with common base audio isolation stage

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a continuation of application Ser. No. 09/608,743, filed on Jun. 30, 2000 which is herein incorporated by reference. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to the field of telecommunications. In particular, this invention is drawn to subscriber line interface circuitry. 
     BACKGROUND OF THE INVENTION 
     Subscriber line interface circuits are typically found in the central office exchange of a telecommunications network. A subscriber line interface circuit (SLIC) provides a communications interface between the digital switching network of a central office and an analog subscriber line. The analog subscriber line connects to a subscriber station or telephone instrument at a location remote from the central office exchange. 
     The analog subscriber line and subscriber equipment form a subscriber loop. The interface requirements of an SLIC result in the need to provide relatively high voltages and currents for control signaling with respect to the subscriber equipment on the subscriber loop. Voiceband communications are low voltage analog signals on the subscriber loop. Thus the SLIC must detect and transform low voltage analog signals into digital data for transmitting communications received from the subscriber equipment to the digital network. For bi-directional communication, the SLIC must also transform digital data received from the digital network into low voltage analog signals for transmission on the subscriber loop to the subscriber equipment. Strict gain and longitudinal balance control are required for subscriber loop applications. 
     In order to meet the strict requirements, high precision high voltage amplifiers are frequently used for processing voiceband signals. The voiceband output signal may be driven as a voltage through a desired output impedance to the subscriber equipment. Alternatively, the tip and ring lines of the subscriber loop are driven by currents corresponding to the voiceband signal while the desired output impedance is synthesized. Disadvantages of high precision high voltage amplifier solutions include the added cost and board area requirements for the SLIC. 
     SUMMARY OF THE INVENTION 
     In view of limitations of known systems and methods, a subscriber line interface circuit is described. A method of coupling an outgoing audio signal to the subscriber line includes the step of receiving the outgoing analog audio signal. The outgoing analog audio signal corresponds to a transformed digital audio signal originating from a digital switching network. The method includes the step of coupling the outgoing audio signal to the subscriber line through a plurality of transistors coupled in a common base configuration. In one embodiment, the method includes the step of receiving linefeed driver control signals on the same signal lines as the outgoing audio signal. The linefeed driver control signals control battery feed to the subscriber line. 
     In one embodiment, a subscriber line interface circuit apparatus includes a first circuit for coupling a received outgoing audio signal to a subscriber line. The first circuit couples the received outgoing audio signal to the subscriber line through a common base isolation stage. In various embodiments, the common base isolation stage comprises a plurality of bipolar junction transistors coupled in a common base configuration or a plurality of field effect transistors coupled in a common gate configuration. 
     In one embodiment a subscriber line interface circuit includes a signal processor and a linefeed driver coupled to receive an outgoing audio signal from the signal processor. The linefeed driver couples the outgoing audio signal to the subscriber line through a common base isolation stage. In one embodiment, the common base isolation stage comprises a plurality of bipolar junction transistors coupled in a common base configuration. In an alternative embodiment, the common base isolation stage comprises a plurality of field effect transistors coupled in a common gate configuration. In various embodiments, the signal processor provides linefeed control signals to the linefeed driver on the same signal lines as the outgoing audio signals. The linefeed control signals control battery feed to the subscriber line. 
     Other features and advantages of the present invention will be apparent from the accompanying drawings and from the detailed description that follows. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present invention is illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like references indicate similar elements and in which: 
         FIG. 1  illustrates one embodiment of a central office exchange including a subscriber line interface circuit (SLIC) coupling subscriber equipment to a digital switching system. 
         FIG. 2  illustrates a block diagram of an SLIC including a signal processor and a linefeed driver. 
         FIG. 3A  illustrates a circuit for driving an outgoing audio signal as a voltage through an output impedance. 
         FIG. 3B  illustrates a circuit for driving the audio output signal as a current through a synthesized output impedance. 
         FIG. 4  illustrates one embodiment of a linefeed driver circuit. 
         FIG. 5A  illustrates the signal processor linefeed control providing an outgoing audio signal to a power circuitry portion of the linefeed driver circuitry. 
         FIG. 5B  illustrates a simplified model of transmission of the outgoing audio signal from the linefeed control portion of the signal processor onto the tip and ring nodes of the subscriber line through a common base isolation stage. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  illustrates functional elements of one embodiment of a subscriber line interface circuit (SLIC)  110  typically associated with plain old telephone services (POTS) telephone lines. The subscriber line interface circuit (SLIC) provides an interface between a digital switching network  120  of a local telephone company central exchange and a subscriber loop  132  including subscriber equipment  130 . 
     The subscriber loop  132  is typically used for communicating analog data signals (e.g., voiceband communications) as well as subscriber loop “handshaking” or control signals. The analog data signals are typically on the order of 1 volt peak-to-peak (i.e., “small signal”). The subscriber loop control signals typically consist of a 48 VDC offset and an AC signal of 40-140 Vrms (i.e., “large signal”). The subscriber loop state is often specified in terms of the tip  180  and ring  190  portions of the subscriber loop. 
     The SLIC is expected to perform a number of functions often collectively referred to as the BORSCHT requirements. BORSCHT is an acronym for “battery feed,” “overvoltage protection,” “ring,” “supervision,” “codec,” “hybrid,” and “test.” 
     Recent transformerless SLIC designs tend to distribute the functional requirements between two integrated circuits based on whether the functions are traditionally associated with the high voltage subscriber loop controls or the low voltage data processing. For example, in one embodiment, the codec is implemented in a low voltage integrated circuit and the remaining functions (e.g., supervision) are implemented primarily in a high voltage integrated circuit such as a bipolar integrated circuit. Although this design tends to offer considerable space, weight, and power efficiencies over designs requiring passive inductive components, this distribution of the functional requirements tends to result in a relatively expensive high voltage integrated circuit. 
       FIG. 2  illustrates one embodiment of a SLIC wherein the BORSCHT functions are distributed between a signal processor  210  and a linefeed driver  220 . Signal processor  210  is responsible for at least the ring control, supervision, codec, and hybrid functions. Signal processor  210  controls and interprets the large signal subscriber loop control signals as well as handling the small signal analog voiceband signals and the digital voiceband data. In one embodiment, the signal processor  210  is an integrated circuit. 
     In one embodiment, the signal processor includes a processor interface  214  to enable programmatic control of the signal processor  210 . The processor interface effectively enables programmatic or dynamic control of battery control, battery feed state control, voiceband signal amplification and level shifting, longitudinal balance, ringing currents, and other subscriber loop control parameters as well as setting thresholds such as a ring trip detection thresholds and an off-hook detection threshold. 
     Signal processor  210  includes a codec for bi-directional transformation of the voiceband communications between the digital and analog domains as is well known in the art. The digital voiceband data is received from the digital switching network on interface  216 . Within the signal processor, the digital voiceband data is coupled to a digital codec interface. An analog codec interface provides outgoing analog voiceband signals to the linefeed driver. The analog codec interface also receives incoming analog voiceband signals from the linefeed driver. The terms “incoming” and “outgoing” used in reference to the voiceband (i.e., audio) signal refer to the intended data flow from the perspective of the digital switching network. Thus, incoming voiceband signals received from the subscriber line are transformed from analog to digital form and provided to the digital switching network. Outgoing voiceband signals from the digital switching network are transformed from digital to analog form and provided to the subscriber line for use by the subscriber equipment. 
     Signal processor  210  receives subscriber line state information from linefeed driver  220  as indicated by tip/ring sense  222 . This information is used to generate control signals for linefeed driver  220  as indicated by linefeed driver control  212 . In one embodiment, the linefeed driver control and outgoing analog voiceband signals are communicated on the same signal lines  212 . Incoming analog voiceband signals are received by the signal processor on line  230 . 
     Linefeed driver  220  maintains responsibility for battery feed to tip  280  and ring  290 . Overvoltage protection is not explicitly illustrated, however, overvoltage protection can be provided by fuses incorporated into linefeed driver  220 , if desired. Linefeed driver  220  includes sense circuitry to provide signal processor  210  with pre-determined sensed subscriber loop operating parameters as indicated by tip/ring sense  222 . Signal processor  210  performs any necessary processing on the sensed parameters in order to determine the operational state of the subscriber loop. For example, differences or sums of sensed voltages and currents are performed as necessary by signal processor  210  rather than linefeed driver  220 . Thus common mode and differential mode components (e.g., voltage and current) of the subscriber loop are calculated by the signal processor rather than the linefeed driver. 
     Linefeed driver  220  modifies the large signal tip and ring operating conditions in response to linefeed driver control  212  provided by signal processor  210 . This arrangement enables the signal processor to perform processing as needed to handle the majority of the BORSCHT functions. For example, the supervisory functions of ring trip, ground key, and off-hook detection can be determined by signal processor  210  based on operating parameters provided by tip/ring sense  222 . 
     Due to strict gain control and longitudinal balance requirements, a precise means of coupling voiceband signals to tip and ring is necessary.  FIG. 3A  illustrates one embodiment of the a.c. signal components associated with the high precision high voltage amplifier providing the outgoing voiceband (i.e., audio) signal to tip and ring. The outgoing audio signal modeled as audio source  310  is driven as a voltage through a desired output impedance R O  using precision high voltage amplifier  320 .  FIG. 3B  illustrates an alternate approach. An outgoing audio signal  350  is driven as a current through a synthesized output impedance using precision high voltage transconductance amplifier  360  and differential error amplifier  370 . The precision amplifiers are typically part of a high voltage integrated circuit which has disadvantages in terms of cost and board area required for implementation. 
       FIG. 4  illustrates one embodiment of a SLIC linefeed driver  410 . In one embodiment, the linefeed driver  410  is implemented as a number of discrete components. Linefeed driver  410  includes a voiceband sensing circuitry  420 , line sensing circuitry  430 , and power circuitry  440 . 
     Voiceband circuitry  420  enables signals corresponding to voiceband communications to be retrieved from the subscriber loop. Nodes  424  and  428  serve to communicate voiceband signals from the subscriber loop to signal processor  210  (i.e., “incoming audio”). Capacitors CR and CT effectively provide AC coupling for the incoming audio signal from the subscriber loop to the signal processor while decoupling signal processor  210  from the DC offsets of the tip  480  and ring  490  nodes. Thus capacitors CR and CT effectively provide DC isolation of the incoming analog audio interface formed by nodes  424  and  428  from the subscriber loop. In the embodiment illustrated, voiceband circuitry  420  provides AC coupling of the incoming analog audio signal between the subscriber loop and the signal processor using only passive components. 
     Line sensing circuitry  430  enables signal processor  210  to determine the tip  480  and ring  490  node voltages as well as the subscriber loop current using sensing resistors RS 1 , RS 2 , RS 3 , and RS 4 . Resistors RT and RR are used to generate a voltage drop for determining the tip and ring currents. In one embodiment, line sensing circuitry  430  consists only of passive discrete components. 
     Referring to  FIG. 2 , tip/ring sense  222  includes a sensed tip signal and a sensed ring signal. In one embodiment, the sensed tip signal includes first and second sensed tip voltages. Resistors RS 1  and RS 2  are used to sense the tip line voltage at each end of RT. Resistors RS 1  and RS 2  convert the sensed tip line voltages to currents suitable for handling by signal processor  210  at nodes  432  and  434 . The difference between the first and second sensed tip voltages is proportional to the tip current. Likewise, the sensed ring signal includes first and second sensed ring voltages. Resistors RS 3  and RS 4  similarly convert sensed ring line voltages at both ends of RR to currents suitable for handling by signal processor  210  at nodes  436  and  438 . The difference between the first and second sensed ring voltages is proportional to the ring current. These calculations, however, can be performed as necessary by the signal processor  210  rather than the linefeed driver  220  circuitry. In addition, these sensed parameters enable the signal processor  210  to determine the subscriber loop voltage and the subscriber loop common mode and differential mode currents. 
     Power circuitry  440  provides the battery feed and other relatively high voltage functions to the subscriber loop in accordance with analog linefeed control signals provided by the signal processor  210  at nodes  442 ,  444 ,  446 , and  448 . Processing of the sensed parameters of the tip and ring lines for generating the linefeed control signals is handled exclusively by signal processor  210 . 
     The subscriber loop current and the tip and ring voltages are controlled by transistors Q 1 -Q 6 . In one embodiment, Q 1 -Q 4  are PNP bipolar junction transistors and Q 5 -Q 6  are NPN bipolar junction transistors. Given that the base terminals of Q 1 -Q 4  are coupled to ground, nodes  442 - 448  need only be approximately 0.7 volts to turn on transistors Q 1 -Q 4 . Due to the small voltage drop between the base and emitters of Q 1 -Q 4 , control of the linefeed circuitry requires relatively low power and thus linefeed driver control currents I 1 -I 4  may be provided by a signal processor  210  implemented as a low voltage complementary metal oxide semiconductor (CMOS) integrated circuit. 
     Transistors Q 1 , Q 4 , and Q 6  (and resistor R 2 ) control the tip voltage  480 . The tip voltage is increased by the application of control current I 1  to Q 1 . The tip voltage (node  480 ) is decreased by the application of control current I 4  to Q 4 . Thus control currents I 1  and I 4  effectively provide a tip control signal for manipulating the tip voltage at node  480 . 
     Similarly, transistors Q 2 , Q 3 , and Q 5  (and resistor R 1 ) control the ring voltage  490 . The application of control current I 3  to Q 3  increases the ring voltage. The ring voltage is decreased by the application of control current I 2  to Q 2 . Control currents I 2  and I 3  effectively provide a ring control signal for manipulating the ring voltage at node  490 . 
     Control currents I 1 -I 4  thus effectively control the large signal subscriber loop current and tip and ring voltages. For example, the ringing signal can be generated by using the control signals at nodes  442 - 448  to periodically reverse the polarity of tip  480  with respect to ring  490  (i.e., battery polarity reversal) at the nominal ringing frequency. 
     Line sensing portion  430  enables signal processor  210  to determine the large signal state of the subscriber loop without the need for intervening active circuitry or level shifters. In one embodiment, line sensing portion  430  comprises only passive discrete components. The linefeed control inputs  442 - 448  enable signal processor  210  to actively manage the large signal state of the subscriber loop. In particular, the large signal AC and DC components of the subscriber loop control protocol can now be controlled directly by a low voltage integrated circuit. The large signal AC and DC control loops are effectively terminated at the signal processor  210 . 
     In other words, the large signal AC and DC control loops are terminated at the low voltage integrated circuit. Thus signal processing and state determination such as off-hook, ring trip, and ring control formerly associated with high power analog circuitry can be handled predominately by a low voltage integrated circuit. In addition, the integrated circuit signal processor can handle processing of the small signal analog voiceband signals from the subscriber loop without the need for intervening active elements or level shifting circuitry. 
     In one embodiment, the outgoing analog audio signal is superimposed on the control currents I 1  and I 3  for power circuitry  440 . Thus the audio signal and the linefeed control signals are provided on the same signal lines to the linefeed driver circuitry. The outgoing audio signal is communicated using nodes  442  and  446 . One advantage of this configuration is that the termination impedance can be set by controlling currents I 1  and I 3 . The use of a programmable signal processor effectively places the value of the termination impedance under programmatic control. 
       FIG. 5A  illustrates the linefeed control  520  portion of the signal processor  210  and the power circuitry  540  of the linefeed driver  410 . Linefeed control  520  includes current sources  522 ,  524 ,  526 , and  528  within the signal processor. The large signal components of currents I 1 -I 4  are controlled by linefeed control  520 . Transistors Q 1  and Q 3  receive the outgoing audio signal from the audio signal current source  530 . Audio signal current source  530  provides a current (I OUT ) corresponding to the outgoing analog audio signal received from the codec. The outgoing analog audio signal originated as a digital audio signal from the digital network before being transformed by the codec. 
     During normal operation, a selected one of transistors Q 5  and Q 6  is “on.” Resistor R L    560  represents the subscriber line impedance load. When transistor Q 5  is on, Q 2 , Q 5 , and R 1  effectively form a DC current source. Alternatively, when transistor Q 6  is on, Q 4 , Q 6 , and R 2  effectively form a DC current source.  FIG. 5B  illustrates a model of  FIG. 5A  when transistor Q 5  is on (normal battery feed). Q 2 , Q 5 , and R 1  are modeled as DC current source  570 . Current sources  522  and  526  are also large signal and thus DC current sources. 
     Transistors Q 1  and Q 3  are coupled in a common base configuration. Transistors Q 1  and Q 3  couple the outgoing audio signal received from the signal processor. The common base isolation stage effectively isolates the signal processor from the DC offset of the tip  580  and ring  590  nodes. Audio current source  530  (I OUT ) manipulates I 1  and I 3  to put the outgoing audio signal onto the tip  580  and ring  590  nodes. 
     A DC bias current is established in Q 1  and Q 3  with non-precision low voltage and high voltage circuitry. The DC bias does not directly affect the audio gain or balance and thus high precision is not required. Subscriber line impedance synthesis can be accomplished by providing sensed tip and ring voltages as feedback for the outgoing audio current source  530 . 
     The gain through the common base stage is the average α (i.e., α AVG ) of Q 1  and Q 2 . For typical transistor betas (β), α approaches 1.0. Thus there is approximately unity gain between I OUT  and the AC component of the current in R L  corresponding to the outgoing audio signal. I OUT  can be increased to accommodate lower transistor gains to maintain a fixed audio amplitude on tip and ring. 
     For example, if each of transistors Q 1  and Q 3  has a β between 100 and 200, the geometric mean of α is calculated as follows: 
     
       
         
           
             α 
             = 
             
               β 
               
                 β 
                 + 
                 1 
               
             
           
         
       
       
         
           
             
               α 
               mean 
             
             = 
             
               
                 
                   
                     α 
                     1 
                   
                    
                   
                     α 
                     2 
                   
                 
               
               = 
               
                 
                   
                     
                       ( 
                       
                         100 
                         101 
                       
                       ) 
                     
                      
                     
                       ( 
                       
                         200 
                         201 
                       
                       ) 
                     
                   
                 
                 = 
                 0.993 
               
             
           
         
       
     
     Current source  530  can be increased by 
     
       
         
           
             
               1 
               0.993 
             
             = 
             1.007 
           
         
       
     
     in order to give a net gain of 1.0. The worst case gain variation would be calculated as follows: 
     
       
         
           
             ɛ 
             = 
             
               
                 
                   ± 
                   20 
                 
                  
                 
                   log 
                    
                   
                     ( 
                     
                       
                         α 
                         
                           m 
                            
                           
                               
                           
                            
                           i 
                            
                           
                               
                           
                            
                           n 
                         
                       
                       
                         α 
                         mean 
                       
                     
                     ) 
                   
                 
               
               = 
               
                 
                   20 
                    
                   
                     log 
                      
                     
                       ( 
                       
                         0.990 
                         0.993 
                       
                       ) 
                     
                   
                 
                 = 
                 
                   
                     ± 
                     0.03 
                   
                    
                   
                       
                   
                    
                   db 
                 
               
             
           
         
       
     
     Thus the worst case gain variation is well within allowable gain variation requirements for subscriber line applications (±0.15 to ±0.50 db). 
     The balance requirement is a measure of the balance between the gains to tip and ring, respectively. This result is calculated as follows: 
     
       
         
           
             balance 
             = 
             
               
                 20 
                  
                 
                   log 
                    
                   
                     ( 
                     
                       
                         
                           α 
                           1 
                         
                         - 
                         
                           α 
                           2 
                         
                       
                       
                         α 
                         mean 
                       
                     
                     ) 
                   
                 
               
               - 
               
                 6.7 
                  
                 
                     
                 
                  
                 db 
               
             
           
         
       
     
     For the example presented above, substitution of the values into the equation for balance yields: 
     
       
         
           
             
               
                 
                   balance 
                   = 
                     
                    
                   
                     
                       ( 
                       
                         
                           0.995 
                           - 
                           0.990 
                         
                         0.993 
                       
                       ) 
                     
                     - 
                     6.7 
                   
                 
               
             
             
               
                 
                   = 
                     
                    
                   
                     
                       - 
                       53 
                     
                      
                     
                         
                     
                      
                     db 
                   
                 
               
             
           
         
       
     
     Greater gain and balance control can be achieved through the use of transistors with higher or better matched betas. Alternatively, other configurations such as Darlington pairs can be used to achieve a greater β. Different types of transistors such as metal oxide semiconductor or junction field effect transistors (i.e., MOSFET or JFET) can be used. The term “common base” includes “common gate” equivalents for MOSFET and JFET transistors. Thus a “common base isolation stage” is intended to include field effect transistors coupled in a common gate configuration. 
     In the preceding detailed description, the invention is described with reference to specific exemplary embodiments thereof. Various modifications and changes may be made thereto without departing from the broader spirit and scope of the invention as set forth in the claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.