Patent Publication Number: US-8111788-B2

Title: Apparatus for estimating and correcting baseband frequency error in a receiver

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 12/688,242, filed Jan. 1, 2010, which is a continuation-in-part of U.S. patent application Ser. No. 12/265,929, filed Nov. 6, 2008, which is a continuation of U.S. patent application Ser. No. 11/209,097, filed Aug. 22, 2005, now U.S. Pat. No. 7,457,347, issued Nov. 25, 2008, which claims the benefit of U.S. Provisional Patent Application No. 60/625,874, filed Nov. 8, 2004; and is a continuation-in-part of U.S. patent application Ser. No. 12/512,203, filed Jul. 30, 2009, which is a continuation of U.S. patent application Ser. No. 11/265,373, filed Nov. 2, 2005, now U.S. Pat. No. 7,570,690, issued Aug. 4, 2009, which claims the benefit of U.S. Provisional Patent Application No. 60/625,188, filed Nov. 5, 2004, which are incorporated by reference as if fully set forth. 
    
    
     FIELD OF INVENTION 
     The present invention is related to wireless receivers. More particularly, the present invention is related to a method and apparatus for estimating and correcting frequency error at baseband in a receiver. 
     BACKGROUND 
     Adaptive receivers, such as a normalized least mean square (NLMS) equalizer used in wireless transmit/receive units (WTRUs) and base stations, optimize their associated filter tap values through an iterative procedure that requires multiple iterations to near convergence. The tap values converge as time passes to a minimum mean square error (MMSE) solution used to perform channel estimation. 
     An NLMS receiver includes an equalizer having an equalizer filter which is continually in the process of converging as it tries to track a time-varying channel. The more complex it is to track the channel, the further the tap values of the equalizer will be from convergence. Generally, faster channels (i.e., channel states that evolve rapidly) are difficult for the equalizer to track. Residual automatic frequency control (AFC) errors in the baseband input into the equalizer cause channels to appear faster than they really are. The increase in the apparent speed of the channel can only be partially mitigated by increasing the step-size of an NLMS algorithm implemented by the NLMS receiver. The increased step-size allows the equalizer filter to more accurately track “fast” channels, but it also increases errors in the MMSE solution which cause degradation in the performance of the receiver. 
     Receivers that employ channel estimation are also degraded by residual AFC errors. Since the bandwidth of the appropriate equalizer filter used in channel estimation is a function of the apparent speed of the channel, large AFC errors force the use of wide-band filters that do not efficiently suppress noise, thus leading to less accurate channel estimates. A simple solution is desired to suppress the residual AFC errors. 
     SUMMARY 
     The present invention is related to an apparatus for estimating and correcting baseband frequency error in a receiver. In one embodiment, an equalizer performs equalization on a sample data stream and generates filter tap values based on the equalization. An estimated frequency error signal is generated based on at least one of the filter tap values. A rotating phasor is generated based on the estimated frequency error signal. The rotating phasor signal is multiplied with the sample data stream to correct the frequency of the sample data stream. In another embodiment, a channel estimator performs channel estimation and generates Rake receiver finger weights based on at least one of the finger weights. An estimated frequency error signal is generated based on at least one of the finger weights. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       A more detailed understanding of the invention may be had from the following description, given by way of example and to be understood in conjunction with the accompanying drawings wherein: 
         FIG. 1  is a block diagram of an example BFC system including a frequency error estimator for removing residual AFC errors in accordance with one embodiment of the present invention; 
         FIG. 2  is a block diagram of the frequency error estimator of the system  100  of  FIG. 1 ; 
         FIG. 3  is a block diagram of an example BFC system in accordance with another embodiment of the present invention; 
         FIG. 4  is a high level flow diagram of a process for correcting the frequency of a sample data stream in a wireless communication receiver having an equalizer that performs equalization in accordance with one embodiment of the present invention; 
         FIG. 5  is a flow diagram of a process for generating the estimated frequency error signal based on a filter tap value extracted from filter tap values generated by the equalizer used in the process of  FIG. 4 ; 
         FIG. 6  is a flow diagram of a process for generating the estimated frequency error signal based on a plurality of extracted tap values that are averaged in accordance with one embodiment of the present invention; 
         FIG. 7  is a flow diagram of a process for comparing the magnitude of the phase difference signal with the value of the threshold signal to determine whether the estimated frequency error signal should be prevented from being updated in accordance with one embodiment of the present invention; 
         FIG. 8  is a flow diagram of a process for comparing the instantaneous power of the phase difference signal with the value of the threshold signal to determine whether the estimated frequency error signal should be prevented from being updated in accordance with one embodiment of the present invention; 
         FIG. 9  is a high level flow diagram of a process for correcting the frequency of a sample data stream in a wireless communication receiver having a channel estimator that performs channel estimation in accordance with one embodiment of the present invention; and 
         FIG. 10  is a flow diagram of a process for generating the estimated frequency error signal based on a finger weight extracted from Rake receiver finger weights generated by the channel estimator used in the process of  FIG. 9 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The preferred embodiments will be described with reference to the drawing figures where like numerals represent like elements throughout. 
     Hereafter, the terminology “WTRU” includes but is not limited to a user equipment (UE), a mobile station, a laptop, a personal data assistant (PDA), a fixed or mobile subscriber unit, a pager, or any other type of device capable of operating in a wireless environment. When referred to hereafter, the terminology “base station” includes but is not limited to an access point (AP), a Node-B, a site controller or any other type of interfacing device in a wireless environment. 
     The features of the present invention may be incorporated into an integrated circuit (IC) or be configured in a circuit comprising a multitude of interconnecting components. 
     Hereinafter, the present invention will be described in terms of the NLMS equalizer. However, it should be noted that the NLMS equalizer based receiver is provided as an example and the present invention can be applied to receivers implementing any other adaptive equalization algorithm and to receivers employing channel estimation such as block based equalizers and rake receivers. 
       FIG. 1  is a block diagram of an exemplary BFC system  100  for removing residual AFC errors in accordance with one embodiment of the present invention. The BFC system  100  may be incorporated in a WTRU or a base station. The BFC system  100  includes a multiplier  102 , an equalizer  104 , a frequency error estimator  106 , a controller  108  and a numerically controlled oscillator (NCO)  110 . The equalizer  104  processes a sample data stream  112  provided by a receiver front end (not shown) via the multiplier  102 . The equalizer may operate in accordance with an NLMS algorithm. However, any other type of adaptive equalizer algorithm may be applied. 
     Filter tap values  114  generated by the equalizer  104  are provided as an input to the frequency error estimator  106 . The frequency error estimator  106  generates an estimated frequency error signal  116 . The residual frequency errors after AFC can be greatly reduced by BFC based solely on observation of at least one tap value in the equalizer  104 , or alternatively from partial channel estimates, such as a Rake finger complex weight estimation. BFC is accomplished by estimating the frequency error based on observation of the one or more taps in the equalizer  104 , generating a correction signal consisting of a complex sinusoid (or rotating phasor), correcting the input samples data stream by multiplying it by the phasor and applying frequency corrected samples  118  to the input of the equalizer  104  in a closed loop fashion. 
     The residual frequency error is estimated by periodically measuring the phase change of one or more of the tap values of the equalizer  104  (or alternatively, partial channel estimates). Much of the phase change measured on the equalizer filter taps  114  from sample to sample is due to noise and fading. However, phase changes due to fading and noise are zero mean (e.g., have a mean value of zero). Therefore, the expected value of any sample average will be zero, i.e., the average value of the signal is zero. Thus, filtering can be used to remove noise and fading components which cause phase change from the overall phase changes (due to, e.g., residual AFC errors) and to recover the slowly varying phase change due to the frequency error. 
     Once the frequency error is estimated by the frequency error estimator  106 , the controller  108  processes the estimated frequency error signal  116  to generate a frequency adjustment signal  120 . The controller  108  may simply adjust the gain of the estimated frequency error signal  116  or may process the estimated frequency error signal  116  with a more complicated algorithm (e.g., a proportional-integral-derivative (PID)). The frequency adjustment signal  120  is fed to the NCO  110  which generates a rotating phasor  122  which corresponds to the frequency adjustment signal  120 . The multiplier  102  multiplies the rotating phasor  122  with the sample data stream  112  to generate the frequency corrected samples  118  input into the equalizer  104 . 
     Residual AFC errors manifest themselves in the baseband as a multiplicative error in the baseband signal and has the form of a complex sinusoid, such as g(t)*exp(j*2pi*f*t) where g(t) is the desired uncorrupted baseband signal and exp(j*2pi*f*t) is the complex sinusoid representing the error. By multiplying by exp(−j*2pi*f*t), the complex sinusoids cancel leaving only the desired signal g(t). The estimated frequency error signal  116  is input to the controller  108  which, in turn, outputs a signal  120  which may be, for example, a scaled (i.e., proportional) version of the input, e.g., four times the value of the estimated frequency error signal  116 . The output signal  120  of the controller  108  may also include other terms such as a term proportional to the integrals and/or derivatives of the estimated frequency error signal  116 . More generally, the output signal  120  could also be clipped to be within some range or have other such non-linear function applied to it. The NCO ( 110 ) takes as an input a frequency value and outputs a constant magnitude complex signal with instantaneous frequency equal to the value of the input, e.g., exp(j*2pi*f*t), where f is the input frequency. 
       FIG. 2  is a block diagram of the frequency error estimator  106  of the BFC system  100  shown in  FIG. 1 . The frequency error estimator  106  includes a tap extraction unit  202 , a delay unit  204 , a conjugate generator  206 , multipliers  208 ,  210 , an arctangent unit  212 , a magnitude detector  214 , an averaging filter  216 , a phase change filter  218  and a comparator  220 . The equalizer generates filter taps  114  which are supplied to the frequency error estimator  106 . 
     In the frequency error estimator  106 , the tap extraction unit  202  extracts and outputs an appropriate tap value or average of tap values onto an output signal  203  from the filter taps  114 , (or alternatively from a channel estimator), to use for performing frequency estimation. For example, at least one appropriate tap value corresponding to an FSP in a particular channel may be extracted from the equalizer filter taps  114 . The tap extraction unit  202  may also track frequency drifting of the extracted tap value. 
     The extracted tap value  203  is forwarded to a delay unit  204  and a conjugate unit  206 . The delay unit  204  delays the extracted tap value  203  for a predetermined period of time by outputting a delayed tap value  205 . The conjugate generator is used to generate a conjugate  207  of the extracted tap value  203 . The multiplier  208  multiplies the delayed tap value  205  by the conjugate tap value  207 . The output  209  of the multiplier  208  has a phase value equal to the phase difference between the delayed tap value  205  and the conjugate tap value  207 . This phase value is proportional to the average frequency of the signal  203  and therefore of the sample data stream  112 . 
     The arctangent unit  212  measures an angle value  213  of the output  209  of the multiplier  208 . The angle value  213  is equal to the phase difference between signal  205  and signal  207 . Averaging the angle value  213  is therefore equivalent to averaging the phase difference between signal  205  and signal  207 . The angle value  213  is filtered by the phase change filter  218  for averaging the angle value  213 . The measured average phase difference and the known delay are used to generate the estimated frequency error signal  116 . 
     For example, with a delay D (sec) and phase measured in radians, the gain of the frequency error estimator  106  is 1/(2*PI*D). The “gain” refers to the conversion of a signal with a net frequency error, (as indicated by signal  114 ), to an observed value of the estimated frequency error signal  116 . If the signal  114  has an average frequency of 1 Hz, then the output value on the estimated frequency error signal  116  will be 1/(2*PI*D). 
     The magnitude detection unit  214  calculates the magnitude of the output  209  of the multiplier  208  and sends a calculated magnitude value  215  to a first input, X, of the comparator  220  and to the averaging filter  216  for averaging. The multiplier  210  multiplies the output signal  217  of the averaging filter  216  (i.e., the average value of signal  215 ) with a threshold factor value  219  (e.g., a scaling factor having a value T) to generate a threshold signal  222  which is sent to a second input, Y, of the comparator  220 . The value of the threshold signal  222  may be set to a fraction of the average amplitude of the output  209  of the multiplier  208 . The threshold factor value, T, may be set, for example, to ⅓. The comparator  220  compares the calculated magnitude value  215  with the value of the threshold signal  222  and sends a hold signal  221  to the phase change filter  218  if the calculated magnitude value  215  is below the value of the threshold signal  222 . 
     The magnitude of the output  209  of the multiplier  208  may be measured and compared to the average amplitude of the output  209  of the multiplier  208 , whereby the phase change filter  218  is paused whenever the magnitude of the output  209  of the multiplier  208  drops below a threshold. When the filter  218  is paused, the estimated frequency error signal  116  does not change (i.e., the signal  116  is not updated), the input  213  is not used, and the internal state of the filter  218  does not change. The hold signal  221  is true whenever the signal  209  is relatively small. This has the effect of discarding the angle values on signal  213  whenever they are noisiest, and improving the estimated frequency error signal  116  when the channel undergoes deep fades. 
     Alternately, a power detector (not shown) may be substituted for the magnitude detector  214  to calculate the average power (i.e., the squared magnitude) of the output  209  of the multiplier  208 , whereby the instantaneous power of the output  209  is compared to some fraction of the average power. Other variations are also possible. 
       FIG. 3  is a block diagram of a BFC system  300  for removing residual AFC errors in accordance with another embodiment of the present invention. The BFC system  100  may be incorporated in a WTRU or a base station. The BFC system  300  includes a multiplier  302 , a rake combiner (or a block equalizer)  304 , a channel estimator  306 , a frequency error estimator  308 , a controller  310 , and an NCO  312 . 
     In a Rake receiver, a finger weight is determined based on the channel estimation on a particular multipath component assigned to a Rake finger. The channel estimator  306  generates Rake receiver finger weights  316  which are provided as an input to the frequency error estimator  308 . The frequency error estimator  308  operates in a fashion similar to the frequency error estimator  106  shown in  FIGS. 1 and 2 . 
     Once the frequency error is estimated by the frequency error estimator  308 , the controller  310  processes the estimated frequency error signal  318  to generate a frequency adjustment signal  322 . The frequency adjustment signal  322  is fed to the NCO  312  which generates a rotating phasor  324  which corresponds to the frequency adjustment signal  322 . The multiplier  302  multiplies the rotating phasor  324  with the sample data stream  314  to generate the frequency corrected samples  320  input into the channel estimator  306  and the Rake combiner  304 . Alternatively, a block equalizer may be used instead of the Rake combiner  304 . 
     The estimated frequency error signal  318  generated by the frequency error estimator  308  is processed by the controller  310  and the NCO  312  which applies a rotating phasor  324  to the sample data stream  314 . 
       FIG. 4  is a high level flow diagram of a process  400  including method steps for correcting the frequency of a sample data stream in a wireless communication receiver having an equalizer that performs equalization in accordance with one embodiment of the present invention. In step  405 , equalization is performed on a sample data stream. In step  410 , filter tap values are generated based on the equalization. In step  415 , an estimated frequency error signal is generated based on at least one of the filter tap values. In step  420 , a rotating phasor signal is generated based on the estimated frequency error signal. In step  425 , the rotating phasor signal is multiplied with the sample data stream to correct the frequency of the sample data stream. 
       FIG. 5  is a flow diagram of a process  500  including method steps for generating the estimated frequency error signal based on a filter tap value extracted from filter tap values generated by the equalizer used in process  400  of  FIG. 4 . In step  505 , an appropriate tap value is extracted from the filter tap values (generated in step  410  of  FIG. 4 ). In step  510 , the extracted tap value is delayed. In step  515 , a conjugate of the extracted tap value is generated. In step  520 , the conjugate of the extracted tap value is multiplied with the delayed extracted tap value to generate a phase difference signal which represents the phase difference between the conjugate extracted tap value and the delayed extracted tap value. In step  525 , the phase difference between the conjugate extracted tap value and the delayed extracted tap value is measured. In step  530 , the estimated frequency error signal is generated by averaging the measured phase difference. In step  535 , the estimated frequency error signal is selectively prevented from being updated based on a value of a threshold signal. 
       FIG. 6  is a flow diagram of a process  600  including method steps for generating the estimated frequency error signal based on a plurality of extracted tap values that are averaged in accordance with one embodiment of the present invention. In step  605 , a plurality of tap values are extracted from the filter tap values (generated in step  410  of  FIG. 4 ). In step  610 , the extracted tap values are averaged to generate an average value of the extracted tap values. In step  615 , the average tap value is delayed. In step  620 , a conjugate of the average tap value is generated. In step  625 , the conjugate of the average tap value is multiplied with the delayed average tap value to generate a phase difference signal which represents the phase difference between the conjugate average tap value and the delayed average tap value. In step  630 , the phase difference between the conjugate average tap value and the delayed average tap value is measured. In step  635 , the estimated frequency error signal is generated by averaging the measured phase difference. In step  640 , the estimated frequency error signal is selectively prevented from being updated based on a value of a threshold signal. 
       FIG. 7  is a flow diagram of a process  700  including method steps for comparing the magnitude of the phase difference signal with the value of the threshold signal to determine whether the estimated frequency error signal should be prevented from being updated in accordance with one embodiment of the present invention. In step  705 , a magnitude of the phase difference signal is calculated. In step  710 , the magnitude of the phase difference signal is averaged. In step  715 , the threshold signal is generated by multiplying a scaling factor with the averaged magnitude of the phase difference signal. In step  720 , the magnitude of the phase difference signal is compared with the value of the threshold signal. In step  725 , the estimated frequency error signal is prevented from being updated if the magnitude of the phase difference signal is below the value of the threshold signal. 
       FIG. 8  is a flow diagram of a process  800  including method steps for comparing the instantaneous power of the phase difference signal with the value of the threshold signal to determine whether the estimated frequency error signal should be prevented from being updated in accordance with one embodiment of the present invention. In step  805 , the instantaneous power of the phase difference signal is calculated. In step  810 , the instantaneous power of the phase difference signal is averaged. In step  815 , the threshold signal is generated by multiplying a scaling factor with the averaged instantaneous power of the phase difference signal. In step  820 , the instantaneous power of the phase difference signal is compared with the value of the threshold signal. In step  825 , the estimated frequency error signal is prevented from being updated if the instantaneous power of the phase difference signal is below the value of the threshold signal. 
       FIG. 9  is a high level flow diagram of a process  900  including method steps for correcting the frequency of a sample data stream in a wireless communication receiver having a channel estimator that performs channel estimation in accordance with one embodiment of the present invention. In step  905 , channel estimation is performed on a sample data stream. In step  910 , Rake receiver finger weights are generated based on the channel estimation. In step  915 , an estimated frequency error signal is generated based on at least one of the finger weights. In step  920 , a rotating phasor signal is generated based on the estimated frequency error signal. In step  925 , the rotating phasor signal is multiplied with the sample data stream to correct the frequency of the sample data stream. 
       FIG. 10  is a flow diagram of a process  1000  including method steps for generating the estimated frequency error signal based on a finger weight extracted from Rake receiver finger weights generated by the channel estimator used in the process  900  of  FIG. 9 . In step  1005 , an appropriate finger weight is extracted from the Rake receiver finger weights (generated in step  910  of  FIG. 9 ). In step  1010 , the extracted finger weight is delayed. In step  1015 , a conjugate of the extracted finger weight is generated. In step  1020 , the conjugate of the extracted finger weight is multiplied with the delayed extracted finger weight to generate a phase difference signal which represents the phase difference between the conjugate extracted finger weight and the delayed extracted finger weight. In step  1025 , the phase difference between the conjugate extracted finger weight and the delayed extracted finger weight is measured. In step  1030 , the estimated frequency error signal is generated by averaging the measured phase difference. In step  1035 , the estimated frequency error signal is selectively prevented from being updated based on a value of a threshold signal. 
     Although the features and elements of the present invention are described in the preferred embodiments in particular combinations, each feature or element can be used alone without the other features and elements of the preferred embodiments or in various combinations with or without other features and elements of the present invention.