Patent Publication Number: US-10790081-B2

Title: Interleaved converters with integrated magnetics

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation-in-part of U.S. patent application Ser. No. 16/006,117, entitled “INTERLEAVED CONVERTERS WITH INTEGRATED MAGNETICS,” filed Jun. 12, 2018, and claims the benefit of U.S. Provisional Application No. 62/674,553, filed May 21, 2018, the entire disclosures of which are hereby fully incorporated herein by reference. 
    
    
     BACKGROUND 
     Power conversion is related to the conversion of electric power or energy from one form to another. Power conversion can involve converting between alternating current (AC) and direct current (DC) forms of energy, AC to AC forms, DC to DC forms, changing the voltage, current, or frequency of energy, or changing some other aspect of energy from one form to another. In that context, a power converter is an electrical or electro-mechanical device for converting electrical energy. A transformer is one example of a power converter, although more complicated systems, including complex arrangements of diodes, synchronous rectifiers, switching transistors, transformers, and control loops, can be used. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Many aspects of the present disclosure can be better understood with reference to the following drawings. The components in the drawings are not necessarily drawn to scale, with emphasis instead being placed upon clearly illustrating the principles of the disclosure. In the drawings, like reference numerals designate corresponding parts throughout the several views. 
         FIG. 1  illustrates a three-phase interleaved LLC converter with a common Y-node at the primary side according to various embodiments described herein. 
         FIG. 2  illustrates a three-phase interleaved LLC converter with a common Y-node at the secondary side according to various embodiments described herein. 
         FIG. 3  illustrates a three-phase interleaved LLC converter with a delta-connected tank at the primary and a configurable secondary side according to various embodiments described herein. 
         FIG. 4  illustrates a three-phase interleaved LLC converter with primary delta-connected resonant capacitors and a common secondary Y-node according to various embodiments described herein. 
         FIG. 5A  illustrates a three-phase interleaved LLC converter with a delta-connected resonant capacitor network at the primary side and a full-bridge at the secondary side according to various embodiments described herein. 
         FIG. 5B  illustrates a three-phase interleaved LLC converter with a common Y-node at the primary and a full-bridge at the secondary side according to various embodiments described herein. 
         FIG. 6A  illustrates a top view of an example transformer having round core legs from which a leakage inductance can be used as a resonant inductor according to various embodiments described herein. 
         FIG. 6B  illustrates a cross-section view of the example transformer shown in  FIG. 6A  according to various embodiments described herein. 
         FIG. 6C  illustrates a top view of an example transformer having elongated core legs according to various embodiments described herein. 
         FIG. 7  illustrates a simplified wiring diagram for an example transformer, a cross-sectional view of the core of the transformer, and a three-dimensional view of the core of the transformer according to various embodiments described herein. 
         FIG. 8  illustrates an example printed circuit board winding implementation for the transformer shown in  FIG. 7  according to various embodiments described herein. 
         FIG. 9A  illustrates an example three-phase interleaved CLLC converter with delta-connected primary resonant capacitors and full-bridge secondary according to various embodiments described herein. 
         FIG. 9B  illustrates an example three-phase interleaved CLLC converter with a common primary Y-node and full-bridge secondary according to various embodiments described herein. 
         FIG. 10A  illustrates a front cross-section view of a proposed transformer according to various embodiments described herein. 
         FIG. 10B  illustrates a back cross-section view of the proposed transformer shown in  FIG. 10A  according to various embodiments described herein. 
         FIG. 11  illustrates a three-dimensional view of the top core section and the bottom core section of the magnetic core of the transformer shown in  FIGS. 10A and 10B  according to various embodiments described herein. 
         FIG. 12  illustrates a reluctance model of the transformer shown in  FIGS. 10A and 10B  according to various embodiments described herein. 
         FIG. 13  illustrates a generalized reluctance model of the transformer shown in  FIGS. 10A and 10B  according to various embodiments described herein. 
         FIG. 14  illustrates a three-dimensional view of magnetic core sections for another transformer according to various embodiments described herein. 
         FIG. 15  illustrates a reluctance model for a transformer including the magnetic core shown in  FIG. 14  according to various embodiments described herein. 
         FIG. 16  illustrates a top cross-section view of another proposed transformer according to various embodiments described herein. 
         FIG. 17  illustrates a top cross-section view of another proposed transformer according to various embodiments described herein. 
         FIG. 18  illustrates an example interleaved CLLC converter with delta-connected primary resonant capacitors, integrated transformer with shielding layers, and full-bridge secondary according to various embodiments described herein. 
         FIG. 19  illustrates a front cross-section view of a proposed transformer according to various embodiments described herein. 
         FIG. 20  shows the voltage distribution on a secondary winding and a shielding layer for one phase of the transformer shown in  FIG. 19  according to various embodiments described herein. 
         FIG. 21  illustrates an example printed circuit board winding implementation for the transformer shown in  FIG. 19  according to various embodiments described herein. 
         FIG. 22  illustrates the layers of another example printed circuit board winding implementation according to various embodiments described herein. 
         FIG. 23  illustrates an example printed circuit board winding implementation for the transformer shown in  FIG. 22  according to various embodiments described herein. 
     
    
    
     DETAILED DESCRIPTION 
     As noted above, power conversion is related to the conversion of electric power or energy from one form to another. Power conversion can involve converting between alternating current (AC) and direct current (DC) forms of energy, AC to AC forms, DC to DC forms, changing the voltage, current, or frequency of energy, or changing some other aspect of energy from one form to another. In that context, a power converter is an electrical or electro-mechanical device for converting electrical energy. A transformer is one example of a power converter, although more complicated systems, including complex arrangements of diodes, synchronous rectifiers, switching transistors, transformers, and control loops, can be used. 
     In the context of power converters, new types of three-phase interleaved LLC and CLLC resonant converters, with integrated magnetics, are described herein. In various examples, the primary sides of the phases in the converters rely upon a half-bridge configuration and include resonant networks coupled to each other in delta-connected or common Y-node configurations. The secondary sides of the phases can rely upon a full-bridge configurations and are coupled in parallel. 
     In other aspects, the transformers of the three phases in the converters are integrated into one magnetic core. By changing the interleaving structure between the primary and secondary windings in the transformers, resonant inductors of the phases can also be integrated into the same magnetic core. A multi-layer PCB can be used as the windings for the integrated magnetics described herein. 
     A number of representative converters are shown in  FIGS. 1-4 . To start,  FIG. 1  illustrates a three-phase interleaved LLC converter  100  with a common Y-node  110  at the primary side according to various embodiments described herein. The converter  100  in  FIG. 1  is provided as a representative example. Other power converters are shown in  FIGS. 2-4, 5A-5B, and 8A-8B . While the converter  100  includes three interleaved phases, additional (or fewer) phases can be interleaved in other examples. 
     A typical LLC converter can have relatively large input and output ripple currents. An interleaved LLC converter, such as the converter  100 , is designed to reduce such ripple currents. By coupling a number of different phases of an LLC converter through a common node or network (e.g., the common Y-node  110  shown in  FIG. 1 ), the different phases of the interleaved LLC converter can achieve current sharing by simply interleaving the primary driving signals between the different phases. 
     As shown in  FIG. 1 , the converter  100  includes transformers  120 - 122 , respectively, for the three phases of the converter  100 . Each of the transformers  120 - 122  includes a primary and a secondary side. Through the common primary Y-node  110 , the three primary phase legs on the primary side of the power converter  100  are coupled together. This coupling at the common primary Y-node  110  achieves interleaving and current sharing among the three phases of the converter  100 . The input to output ratio of the converter  100  at resonant frequency is: 
     
       
         
           
             
               
                 V 
                 O 
               
               
                 V 
                 IN 
               
             
             = 
             
               1 
               
                 2 
                 ⁢ 
                 n 
               
             
           
         
       
     
       FIG. 2  illustrates a three-phase interleaved LLC converter  200  with a common Y-node  210  at the secondary side according to various embodiments described herein. As shown in  FIG. 2 , the converter  200  includes transformers  220 - 222 , respectively, for the three phases of the converter  200 . Each of the transformers  220 - 222  includes a primary and a secondary side. Through the common secondary Y-node  210 , the three secondary phase legs on the secondary side of the power converter  200  are coupled together. When using the common secondary Y-node  210 , the secondary side is in a half-bridge configuration and behaves similar to a voltage doubler. The input to output ratio of the converter  100  at resonant frequency is: 
     
       
         
           
             
               
                 V 
                 O 
               
               
                 V 
                 IN 
               
             
             = 
             
               1 
               n 
             
           
         
       
     
       FIG. 3  illustrates a three-phase interleaved LLC converter  300  with a delta-connected tank  302  at the primary and a configurable secondary side  304  according to various embodiments described herein. In the delta-connected tank  302 , the resonant tanks of the transformers are coupled in a delta-connection. The primary switching nodes of the three phases of the converter  300  are connected to the three nodes of the delta-connected primary tank  302 , respectively. The configurable secondary side  304  can be configured into a delta-connection or a Y-connection. 
       FIG. 4  illustrates a three-phase interleaved LLC converter  400  with delta-connected resonant capacitors  402  on the primary side and a common secondary Y-node  404  on the secondary side according to various embodiments described herein. The delta-connected resonant capacitors  402  couple the three phases of the converter  400  together on the primary side. The secondary side includes the common secondary Y-node  404 . 
     Similar to the converters  100 ,  200 , and  300 , the primary phase legs in the converter  400  comprise resonant tank circuits (e.g., resonant capacitor networks, LC networks, LLC networks, etc.) used to transfer energy to the secondary side of the converter  400 . For example, the first primary phase leg in the converter  400  includes a resonant tank circuit including the inductor L r1 , the inductor L m1 , and a combination of the capacitors C Δ_13  and C Δ_12 . The inductor L r1  is formed from the leakage inductance of the transformer  410 , and the inductor L m1  is formed from the magnetization inductance of the transformer  410 . Similarly, the inductors L r2  and L r3  and the inductors L m2  and L m3  can be formed from the leakage and magnetization inductances of the transformers of the second and third phase legs of the converter  400 . The inductors L r1 , L r2 , and L r3  and the inductors L m1 , L m2 , and L m3  can be integrated into one magnetic component similar to one or more of those shown in U.S. Patent Application Pub. No. 2016/0254756, the entire contents of which is hereby incorporated herein by reference. 
     Turning to other configurations of LLC converters,  FIG. 5A  illustrates a three-phase interleaved LLC converter  500  with delta-connected resonant capacitor networks at the primary side and a full-bridge at the secondary side. The converter  500  in  FIG. 5A  is provided as a representative example. While the converter  500  includes three interleaved phases, additional (or fewer) phases can be interleaved in other examples. Further, the transformer  530  of the converter  500  can rely upon a turn ratio of n: 1  as an example, but any suitable ratio can be relied upon. 
     As shown in  FIG. 5A , the converter  500  includes three interleaved primary phase legs on a primary side of the converter  500 . Each of the phase legs includes a primary-side resonant tank circuit. The resonant tank circuits of the phase legs on the primary side are electrically coupled to each other in a delta-connected resonant capacitor configuration. The secondary side is full-bridge configuration, and the outputs are taken in parallel. The secondary side includes a first full bridge  521 , a second full bridge  522 , and a third full bridge  523 . 
     The first phase leg of the converter  500  is formed of the synchronous rectifiers  511  and a first primary resonant tank circuit. The first primary resonant tank circuit includes the inductor L r1 , the inductor L m1 , and a combination of the capacitors C Δ_13  and C Δ_12 . The second phase leg is formed of the synchronous rectifiers  512  and a second primary resonant tank circuit. The second primary resonant tank circuit includes the inductor L r2 , the inductor L m2 , and a combination of the capacitors C Δ_12  and C Δ_23 . The third phase leg is formed of the synchronous rectifiers  512  and a third primary resonant tank circuit. The third primary resonant tank circuit includes the inductor L r3 , the inductor L m3 , and a combination of the capacitors C Δ_13  and C Δ_23 . The inductors L r1 , L r2 , and L r3  can be embodied as the leakage inductances from the transformer  530  of converter  500 . The inductors L m1 , L m2 , and L m3  can be embodied as the magnetization inductances from the transformer  530  of converter  500 . As shown in  FIG. 5 , the inductors L r1 , L r2 , and L r3  and the inductors L m1 , L m2 , and L m3  can be integrated together in the transformer  530  having a magnetic core  532 . 
     As another example,  FIG. 5B  illustrates a three-phase interleaved LLC converter  550  with a common Y-node at the primary side and a full-bridge at the secondary side according to various embodiments described herein. The three-phase interleaved LLC converter  550  uses a common Y-node at primary side. The secondary side is full-bridge configuration, and the outputs are taken in parallel. The full-bridge configuration at the secondary side in both the converters  500  and  550  is more suitable for high frequency operation, because the currents in the secondary-side devices are half of that in the half-bridge configuration. Further, the AC current loop in the secondary side is minimized. The transformers in the converters  500  and  550  can be integrated into one magnetic component. 
       FIG. 6A  illustrates a top view of an example transformer  600  according to various embodiments described herein, and  FIG. 6B  illustrates a cross-section view of the example transformer  600 . The transformers in the converters  500  and  550  shown in  FIGS. 5A and 5B  (and the converters  900  and  950  shown in  FIGS. 9A and 9B ) can be embodied, as one example, by the transformer  600 . The transformer  600  includes a magnetic core  610  having three core legs  611 - 613 . Each of the core legs  611 - 613  can be associated with one phase leg of a power converter, such as the power converters  500  and  550 . The core legs  611 - 613  have a round or circular cross-sectional profile, although other cross-sectional profile shapes can be relied upon. For example,  FIG. 6C  illustrates a top view of an example transformer  650  having elongated core legs according to various embodiments described herein. 
     Referring between  FIGS. 6A and 6B , the transformer  600  includes a number of primary-side and secondary-side windings that extend around the core legs  611 - 613 . As best shown in  FIG. 6B , the windings  620 A- 620 D extend around the core leg  611 , the windings  630 A- 630 D extend around the core leg  612 , and the windings  640 A- 640 D extend around the core leg  613 . In the example shown, the windings  620 A and  620 D serve as secondary-side windings, and the windings  620 B and  620 C serve as primary-side windings, although other arrangements are within the scope of the embodiments. The secondary windings  620 A and  620 D can include 2 turns (e.g., one turn in the winding  620 A electrically coupled to one turn in the winding  620 D) around the core leg  613 , and the primary windings  620 B and  620 C can include 12 turns (e.g., six turns in the winding  620 B electrically coupled to six turns in the winding  620 C) around the core leg  613 , for a primary to secondary turns ration of 6:1, as an example, although other turns ratios can be used. The windings  630 A- 630 D and  640 A- 640 D can include similar arrangements and turns of primary and secondary windings. 
     The windings  620 A- 620 D,  630 A- 630 D, and  640 A- 640 D can be embodied as metal (e.g., copper) traces on a multi-layer printed circuit board (PCB) in one embodiment. In that case, the windings  620 A,  630 A, and  630 A can be separated from the windings  620 B,  630 B, and  630 B, and so on, by separating them from each other on different layers of the PCB, as shown in  FIG. 6B . Connections between traces and layers in the PCB can be achieved through the use of plated vias in the PCB, for example, or other suitable means. Additionally, the top windings  620 A,  630 A, and  640 A can also include bonding pads  621 ,  631 , and  641 , respectively, for direct electrical coupling to synchronous rectifiers, inductors, capacitors and other discrete and/or integrated components. The bottom windings  620 D,  630 D, and  640 D can also include similar bonding pads. 
     There is some leakage inductance associated with the transformer  600 . Particularly, there is leakage inductance, L r1 , associated with the windings  620 A- 620 D and the core leg  611 . There is also leakage inductance, L r2 , associated with the windings  630 A- 630 D and the core leg  612 , and leakage inductance, L r3 , associated with the windings  640 A- 640 D and the core leg  613 . Leakage inductance is a property of a transformer that causes the windings of the transformer to appear to have some pure inductance (i.e., leakage inductance) in series with the magnetization inductance of the mutually-coupled primary and secondary windings in the transformer. Leakage inductance is typically an undesirable property of transformers. According to aspects of the embodiments described herein, however, the leakage inductances of the transformer  600  can be relied, in part, for use in the resonant tank circuits of the interleaved phase legs in power converters. As described in further detail below, the leakage inductances in the transformer  600  (and other transformers described herein) can be primarily controlled or based on the design of the windings and the magnetic core used to form the transformer  600 . In the transformer  600 , the leakage inductances, L r1 , L r2 , and L r3  are relatively small and relatively difficult to control or determine. 
     Other transformer structures can be relied upon to create larger, more tailored leakage inductances.  FIG. 7  illustrates a simplified wiring diagram for an example transformer  700 , a cross-sectional view of the core  710  of the transformer  700 , and a three-dimensional view of the core  710  of the transformer  700  according to various embodiments described herein. The transformers in the converters  500  and  550  shown in  FIGS. 5A and 5B  (and the converters  900  and  950  shown in  FIGS. 9A and 9B ), among others, can be embodied by the transformer  700 . The leakage inductances of the transformer  700  can be larger than those of the transformer  600 , for example, based on the design factors described below. The leakage inductances can also be controlled or determined based on the design factors described below. 
     The transformer  700  includes windings and core legs for three phases of a power converter. In  FIG. 7 , the primary winding  720 A and the secondary winding  720 B are windings for the first phase of the power converter. Further, the primary winding  721 A and the secondary winding  721 B are windings for the second phase of the power converter, and the primary winding  722 A and the secondary winding  722 B are windings for the third phase of the power converter. 
     The core  710  includes two core legs for each phase of the power converter. In  FIG. 7 , the core leg  711 A and the core leg  711 B are two core legs for the first phase of the power converter. The core leg  712 A and the core leg  712 B are two core legs for the second phase of the power converter, and the core leg  713 A and the core leg  713 B are two core legs for the third phase of the power converter. Thus, the core  710  includes six core legs in total. 
     The portion of the primary winding  720 A that extends around the core leg  711 B contributes to the leakage inductance for the first phase leg of the transformer  700 . This leakage inductance can be used as part of a resonant tank circuit for a phase leg of a power converter. For example, this leakage inductance can be relied upon as the inductor L r1  in the first phase leg of the converter  500  shown in  FIG. 5A , the converter  550  shown in  FIG. 5A , and other power converters. Similarly, the portion of the primary winding  721 A that extends around the core leg  712 B contributes to the leakage inductance for the second phase leg of the transformer  700 . This leakage inductance can be relied upon as the inductor L r2 , for example, in the second phase leg of the converter  500  shown in  FIG. 5A , the converter  550  shown in  FIG. 5A , and other power converters. Additionally, the portion of the primary winding  722 A that extends around the core leg  713 B contributes to the leakage inductance for the third phase leg of the transformer  700 . This leakage inductance can be relied upon as the inductor L r3 , for example, in the third phase leg of the converter  500  shown in  FIG. 5A , the converter  550  shown in  FIG. 5A , and other power converters. 
     When the transformer  700  is relied upon in a power converter, the transformer  700  forms three resonant inductors (e.g., the leakage inductances formed from the primary windings  720 A- 722 A around the core legs  711 B- 713 B) and three transformers (formed from the primary windings  720 A- 722 A, the secondary windings  720 B- 722 B, and the core legs  711 A- 713 A). Air gaps i g_r  exist between the core legs  711 B- 713 B of the core section  701  and the core section  702 . Air gaps l g_m  also exist between the core legs  711 A- 713 A of the core section  701  and the core section  702 . The leakage inductances L r1 , L r2 , and L r3  of the transformer  700  can be controlled or determined based on the cross-sectional areas (i.e., Δ e_r ) of the core legs  711 B- 713 B and the size of the air gap l g_r . The magnetizing inductances L m1 , L m2 , and L m3  of the transformer  700  can be controlled or determined based on the cross-sectional areas (i.e., Δ e_m ) of the core legs  711 A- 713 A and the size of the air gap l g_m . according to the following expression: 
     
       
         
           
             
               L 
               N 
             
             = 
             
               
                 
                   L 
                   m 
                 
                 
                   L 
                   r 
                 
               
               = 
               
                 
                   
                     A 
                     e_m 
                   
                   / 
                   
                     l 
                     g_m 
                   
                 
                 
                   
                     A 
                     e_r 
                   
                   / 
                   
                     l 
                     g_r 
                   
                 
               
             
           
         
       
     
     The windings of the transformer  700  can be implemented using a number of layers in a PCB, such as the 4-layer PCB winding  800  shown in  FIG. 8 . The 4-layer PCB winding  800  shown in  FIG. 8  can be used for one phase leg of the transformer  700  shown in  FIG. 7 . The top layer  801  and the bottom layer  802  can be coupled in parallel to form one turn of the secondary winding  720 B shown in  FIG. 7 . The middle layers  811  and  812  can be electrically coupled together through the vias  820  to form four turns of the primary winding  720 A. In other examples, PCB windings with more layers can be used to reduce winding conduction loss. 
     Turning to other embodiments,  FIG. 9A  illustrates an example three-phase interleaved CLLC converter  900  with delta-connected primary resonant capacitors and full-bridge secondary according to various embodiments described herein.  FIG. 9B  illustrates an example three-phase interleaved CLLC converter  950  with a common primary Y-node and full-bridge secondary according to various embodiments described herein. 
       FIG. 10A  illustrates a front cross-section view of a proposed transformer  1000 , and  FIG. 10B  illustrates a back front cross-section view of the transformer  1000  shown in  FIG. 10A . Resonant inductors on both the primary and secondary sides of the example converter  900  shown in  FIG. 9A  and the example converter  950  shown in  FIG. 9B  can be realized using the transformer  1000 . 
     As shown in  FIGS. 10A and 10B , the transformer  1000  includes a top core section  1001 , a bottom core section  1002 , and a number of windings. The transformer  1000  includes core legs A 1  and A 2  for a first phase leg of a power converter, core legs B 1  and B 2  for a second phase leg of the power converter, and core legs C 1  and C 2  for a third phase leg of the power converter. Primary and secondary windings are wound around the core legs A 1  and A 2 , although the distribution of the primary and secondary windings is uneven between the core legs A 1  and A 2 . Similarly, primary and secondary windings are wound around the core legs B 1  and B 2 , although the distribution of the windings is uneven between them. Primary and secondary windings are also wound around the core legs C 1  and C 2 , although the distribution of the windings is uneven between them. 
     For the first phase leg, four primary PCB windings  1010 - 1013  are wound around the core leg A 1 , but only two secondary PCB windings  1020  and  1021  are wound around the core leg A 1 . Further, two primary PCB windings  1014  and  1015  are wound around the core leg A 2 , and four secondary PCB windings  1022  and  1025  are wound around the core leg A 2 , for turns ratio of 6:6 among the core legs A 1  and A 2 . This uneven distribution of primary and secondary windings between the core legs A 1  and A 2  is the same around the core legs B 1  and B 2  for the second phase leg and the core legs C 1  and C 2  for the third phase leg. 
       FIG. 11  illustrates a three-dimensional view of the top core section  1001  and the bottom core section  1002  of the magnetic core of the transformer  1000  shown in  FIGS. 10A and 10B . The core legs A 1 , A 2 , B 1 , B 2 , C 1 , and C 2  are shown as being rectangular with rounded corners or ends in  FIG. 11 , but the core legs A 1 , A 2 , B 1 , B 2 , C 1 , and C 2  can be formed in any suitable shape. In the embodiment shown in  FIG. 11 , the top core section  1001  is formed as a single piece, and the bottom core section  1002  is formed as a single piece. Both the top core section  1001  and the bottom core section  1002  can be formed from any suitable material or materials. 
     The reluctance model of the transformer  1000  is shown in  FIG. 12 . With this reluctance model, the transformer equivalent magnetizing inductance and leakage inductance can be calculated for each phase, as follows: 
     
       
         
           
             
               
                 L 
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                 16 
                 
                   R 
                   g 
                 
               
             
             , 
             
               
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     The magnetic structure shown in  FIGS. 10A, 10B, and 11  is not limited to 6:6 turns ratio, however. For other turns ratios, a more generalized reluctance model is shown in  FIG. 13 . For the generalized reluctance model, the turns ratio in each phase is Np 1 +Np 2 :Ns 1 +Ns 2 . Np 1  and Ns 1  are the number of primary windings and secondary windings on A 1 , respectively, while Np 2  and Ns 2  are the number of primary windings and secondary windings on A 2 , respectively. With this generalized model, the transformer equivalent magnetizing inductance and leakage inductance can be calculated for each phase, as follows: 
     
       
         
           
             
               
                 L 
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                   4 
                   ⁢ 
                   
                     N 
                     
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                       1 
                     
                   
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     The magnetic structure shown in  FIGS. 10A, 10B, and 11  can also be realized in another way. Instead of using one core with six core legs, two separate cores can be used with three core legs each core, as shown in  FIG. 14 . As shown in  FIG. 14 , a magnetic core includes two separate top core sections,  1001 A and  1001 B. The magnetic core also includes two separate bottom core sections,  1002 A and  1002 B. 
     The corresponding reluctance model for a transformer including the magnetic core  1100  is shown in  FIG. 15 . From the reluctance model, the transformer equivalent magnetizing inductance and leakage inductance can be calculated for each phase, as follows: 
     
       
         
           
             
               
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                       1 
                     
                   
                 
                 . 
               
             
           
         
       
     
     The interleaved LLC converters described herein can be extended to interleaved LLC converter with any odd number of phases. The proposed magnetic structures shown in  FIGS. 6A, 6B, 6C, 7, 8, 10A, 10B, 11, and 14  can be extended to use with any number of layers of PCB windings. 
     Turning to other embodiments,  FIG. 16  illustrates a top cross-section view of another proposed transformer according to various embodiments described herein. The transformer includes three upper core legs A 1 , B 1 , and C 1  and three lower core legs A 2 , B 2 , and C 2 , with a number of primary and secondary windings around each core leg. As examples, the transformer shown in  FIG. 16  can be implemented using the top core section  1001  and the bottom core section  1002  shown in  FIG. 11 . Alternatively, the transformer can be implemented using the two separate top core sections,  1001 A and  1001 B, and the two separate bottom core sections,  1002 A and  1002 B, shown in  FIG. 14 . Any suitable type(s) of windings can be used with the transformer shown in  FIG. 16 . The windings are not limited to litz-wire, solid copper wire, or copper plate, as PCB based windings can also be relied upon as described herein. 
     In  FIG. 16 , the turn direction of the primary and secondary windings for all cores is arranged such that the magnetizing flux is in the same direction for the three upper core legs A 1 , B 1 , and C 1  and the three lower core legs A 2 , B 2 , and C 2  (i.e., out of the page for every core leg). This type of integrated transformer is suitable for delta or Y-node connected primary and secondary side because it lacks the ability to limit third order harmonics. 
     The number of primary and secondary windings of the transformer shown in  FIG. 16  can be distributed evenly between the three upper core legs A 1 , B 1 , and C 1  and the three lower core legs A 2 , B 2 , and C 2 . For example, a 6:6 turns ratio can be relied upon, although other evenly distributed turns ratios can be relied upon. 
     If an additional resonant inductor is needed for use in an LLC or CLLC converter, the number of primary and secondary windings of the transformer shown in  FIG. 16  can be unevenly distributed between the upper core legs A 1 , B 1 , and C 1  as compared to the lower core legs A 2 , B 2 , and C 2 . For example, 4 primary turns and 2 secondary turns can be used on the upper core legs A 1 , B 1 , and C 1 , while 2 primary turns and 4 secondary turns can be used on the lower core legs A 2 , B 2 , and C 2 , for an unevenly distributed 6:6 turns ratio. This is similar to the embodiments described above with reference to  FIGS. 10A and 10B , and the reluctance model is similar to that shown in  FIG. 12 . 
     The transformer shown in  FIG. 16  is not limited to a 6:6 turns ratio for each of the core legs, however. For a more generalized magnetic structure, the turns ratio in each phase can be defined as Np 1 +Np 2 :Ns 1 +Ns 2 , where Np 1  and Ns 1  are the number of primary and secondary windings, respectively, on the core leg A 1 , and Np 2  and Ns 2  are the number of primary and secondary windings, respectively, on the core leg A 2 . The reluctance model for this generalized structure is similar to that shown in  FIG. 13 . Finally, in any of the examples described for  FIG. 16 , the interleaved resonant converters and integrated magnetic structures can be extended to any odd number of phases and to the use of PCB windings with any number of layers. 
       FIG. 17  illustrates a top cross-section view of another proposed transformer according to various embodiments described herein. The transformer shown in  FIG. 17  can also be implemented using the top core section  1001  and the bottom core section  1002  shown in  FIG. 11 . Alternatively, the transformer can be implemented using the two separate top core sections,  1001 A and  1001 B, and the two separate bottom core sections,  1002 A and  1002 B, shown in  FIG. 14 . Any suitable type(s) of windings can be used with the transformer shown in  FIG. 17 . The windings are not limited to litz-wire, solid copper wire, or copper plate, as PCB based windings can also be relied upon as described herein. 
     As compared to the transformer shown in  FIG. 16 , the primary and secondary windings in  FIG. 17  are arranged (i.e., in turn direction) such that the magnetizing flux in the upper core legs A 1 , B 1 , and C 1  is in a different direction than the magnetizing flux in the lower core legs A 2 , B 2 , and C 2 . As shown, the magnetizing flux in the upper core legs A 1 , B 1 , and C 1  is into the page, and the magnetizing flux in the lower core legs A 2 , B 2 , and C 2  is out of the page. This type of integrated transformer can limit third order harmonic currents in the primary and/or second side windings. Additionally, simulations of the transformer shown in  FIG. 17  demonstrate less core loss as compared to the transformer shown in  FIG. 16 . In one simulation, the transformer shown in  FIG. 17  exhibited around 20% less core loss than the transformer shown in  FIG. 16 . 
     The number of primary and secondary windings of the transformer shown in  FIG. 17  can be distributed evenly between the upper core legs A 1 , B 1 , and C 1  and the lower core legs A 2 , B 2 , and C 2 . For example, a 6:6 turns ratio can be relied upon, although other evenly distributed turns ratios can be relied upon. 
     If an additional resonant inductor is needed for use in an LLC or CLLC converter, the number of primary and secondary windings of the transformer shown in  FIG. 17  can be unevenly distributed between the upper core legs A 1 , B 1 , and C 1  as compared to the lower core legs A 2 , B 2 , and C 2 . For example, 4 primary turns and 2 secondary turns can be used on the upper core legs A 1 , B 1 , and C 1 , while 2 primary turns and 4 secondary turns can be used on the lower core legs A 2 , B 2 , and C 2 , for an unevenly distributed 6:6 turns ratio. This is similar to the embodiments described above with reference to  FIGS. 10A and 10B , and the reluctance model is similar to that shown in  FIG. 12 . 
     The transformer shown in  FIG. 17  is not limited to a 6:6 turns ratio for each of the core legs, however. For a more generalized magnetic structure, the turns ratio in each phase can be defined as Np 1 +Np 2 :Ns 1 +Ns 2 , where Np 1  and Ns 1  are the number of primary windings and secondary windings, respectively, on the core leg A 1 , and Np 2  and Ns 2  are the number of primary windings and secondary windings, respectively, on the core leg A 2 . The reluctance model for this generalized structure is similar to that shown in  FIG. 13 . Finally, in any of the examples described for  FIG. 17 , the interleaved resonant converters and integrated magnetic structures can be extended to any odd number of phases and to the use of PCB windings with any number of layers. 
     Turning to other embodiments,  FIG. 18  illustrates an example interleaved CLLC converter  1200  with delta-connected primary resonant capacitors, integrated transformer  1300  with shielding layers  1310 , and full-bridge secondary according to various embodiments described herein. As shown in  FIG. 18 , the shielding layers  1310  are provided between the primary and secondary windings of each phase leg in the transformer  1300 . The shielding layers  1310  are electrically connected to the primary side ground of the converter  1200 . Therefore, common mode (CM) noise current induced by the primary windings in the transformer  1300  flows to the shielding layers  1310  and circulates back to the primary side ground. In one example case, the shielding layers  1310  can be made the same as the secondary windings, both single-turn windings, so they have the same voltage potential distribution. Thus, even if there is a parasitic capacitance between the shielding layers  1310  and the secondary windings, there is no common mode current between them because the voltage potential difference across the parasitic capacitance is zero. 
       FIG. 19  illustrates a front cross-section view of the transformer  1300  shown in  FIG. 18  according to various embodiments described herein. The transformer  1300  includes two shielding layers  1310 A and  1310 B. The first shielding layer  1310 A is placed between a first secondary winding  1320 A and a first primary winding  1330 A. The second shielding layer  1310 B is placed between a second secondary winding  1320 B and a second primary winding  1330 B. The arrangement of the shielding layers between the primary and secondary windings is the same for all core legs as shown. 
       FIG. 20  shows the voltage distribution on a secondary winding and a shielding layer for one phase of the transformer shown in  FIG. 19  according to various embodiments described herein. The primary winding is not shown in  FIG. 20  for simplicity. Two terminals of the secondary windings are marked as A and B, and those of the shielding layers are marked as A′ and B′. The centers of the shielding layers are connected to primary-side ground. The windings can be stretched along the x-axis to map the voltage potential at each point on the windings to the U-x coordinate at the right side of  FIG. 20 . Since the secondary winding and shielding are identical, the voltage potentials of both at the same position on the x-axis are identical, so the two curves on the U-x coordinate overlap each other, and have U=V at x=0 and U=−V at x=L. 
       FIG. 21  illustrates an example PCB winding implementation for the transformer shown in  FIG. 19  according to various embodiments described herein.  FIG. 21  illustrates the windings for one phase for simplicity because the windings for each of the three phases are identical. The shielding layers only need to cover the common area between the primary windings and the secondary windings. 
       FIG. 22  illustrates the layers of another example PCB winding implementation according to various embodiments described herein.  FIG. 22  illustrates a cross-section view of a 12-layer PCB with shielding. Here, the primary and secondary windings are arranged to reduce the number of contact surfaces. As a result, good interleaving is maintained for AC winding loss reduction, and only four shielding layers are needed between the primary and secondary windings to complete shield CM noise. 
       FIG. 23  illustrates an example PCB winding implementation for the transformer shown in  FIG. 22  according to various embodiments described herein. As shown, Layer  2 , Layer  4 , Layer  7 , and Layer  10  are shielding layers and have the exact same layout as the secondary windings next to them. By using the same layout, the voltage potential between the shielding layers and the secondary windings will be the same. In other words, since the layout of the shielding layer is the same as the layout of the adjacent secondary winding, there is no voltage change rate difference and no CM current between them. Additionally, since the shielding layers are electrically connected to the primary-side ground, current between the primary windings and the shielding layers will cycle inside the converter and does not contribute to CM noise. 
     Another unique feature of the layout shown in  FIG. 23  is the half turn concept. For Layer  6 , two half turns compose the whole turn. The benefits is that the total winding length can be reduced because the winding only needs to pass from one post to another and back once using the nearest path. 
     The embodiments described herein include new three-phase interleaved LLC and CLLC resonant converters with integrated magnetic structures. Certain features and advantages include primary side coupling of different phases of LLC or CLLC converters through a delta-connected resonant capacitor network or a common Y-node to achieve automatic current sharing. In some cases, the secondary side can rely upon a full-bridge configuration, and the outputs of different phases on the secondary side can be connected in parallel to minimize the AC current loop. 
     In one magnetic structure, the transformers for three phases of a power converter, for example, can be integrated into one magnetic core with three core legs, and the leakage inductances of each core leg can be used as resonant inductors for the three phases of the power converter. In another example, three inductors and three transformers can be integrated into one magnetic core with six core legs, and the resonant inductances and magnetizing inductances can be controlled independently. 
     In another magnetic structure, the transformers for three phases of a power converter can be integrated into one magnetic core with six core legs (e.g., three top and three bottom core legs), and the leakage inductance of each transformer can be used as a resonant inductor. The windings can be arranged so that the top and bottom core legs have the same flux direction. Alternatively, the windings can be arranged so that the top and bottom core leg have reverse flux directions. Third order harmonics can be suppressed as compared to when the top and bottom core legs have the same flux direction. Also, the core loss can be smaller due to the distributed flux due to the reverse flux directions. [ 86 ] In another magnetic structure, six inductors and three transformers can be integrated into one magnetic core with six core legs. The resonant and magnetizing inductances can be controlled by adjusting an air gap between the cores. The ratio between the resonant inductances and the magnetizing inductances can be changed by changing the primary and secondary winding distributions. The windings can be arranged so that the top and bottom core legs have the same flux direction. Alternatively, the windings can be arranged so that the top and bottom core leg have reverse flux directions. Third order harmonics can be suppressed as compared to when the top and bottom core legs have the same flux direction. Also, the core loss can be smaller due to the distributed flux due to the reverse flux directions. 
     In another magnetic structure, six inductors and three transformers can be integrated into two magnetic cores with three core legs for each core. The resonant and magnetizing inductances can be controlled by adjusting an air gap between the cores. The ratio between the resonant inductances and the magnetizing inductances can be changed by changing the primary and secondary winding distributions. The windings can be arranged so that the top and bottom core legs have the same flux direction. Alternatively, the windings can be arranged so that the top and bottom core leg have reverse flux directions. Third order harmonics can be suppressed as compared to when the top and bottom core legs have the same flux direction. Also, the core loss can be smaller due to the distributed flux due to the reverse flux directions. 
     A multi-layer PCB winding can be employed in any of the transformers described herein, and synchronous rectifiers can be integrated as part of the windings. Shielding layers can also be employed in any of the transformers described herein to block CM noise. 
     The above-described examples of the present disclosure are merely possible examples of implementations set forth for a clear understanding of the principles of the disclosure. Many variations and modifications can be made without departing substantially from the spirit and principles of the disclosure. All such modifications and variations are intended to be included herein within the scope of this disclosure and protected by the following claims.