Patent Publication Number: US-8526522-B2

Title: Single carrier high rate wireless system

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation application of and claims the benefit of priority under 35 U.S.C. §120 from U.S. Pat. No. 8,050,339, filed Mar. 10, 2008, the entire contents of which is incorporated herein by reference. U.S. application Ser. No. 12/045,342 is based upon and claims the benefit of priority under 35 U.S.C. §119 from prior European Patent Application No. 07 006 738.4, filed Mar. 30, 2007. 
    
    
     FIELD OF INVENTION 
     The present invention relates to the field of single carrier wireless communication, in particular to the management of the time frame structure for the single carrier wireless communication. 
     PROBLEM 
     For high rate indoor single carrier wireless systems beyond 1 Gbps, the wireless channel delay spread might be over tens of symbols which makes conventional time-domain channel equalizers including linear, decision feedback or maximum likelihood sequence estimation (MLSE) equalizer unrealistic.
         The adaptive equalizer including either linear or decision feedback equalizer is difficult to converge with short training period, because the required number of equalizer taps increases in order to cover the wireless channel delay spread which is over tens of symbols.   The complexity of maximum likelihood sequence estimation (MLSE) or Viterbi equalizer grows exponentially with the number of symbols included in wireless channel delay spread because the required constraint length of Viterbi algorithm increases when wireless channel delay spread is over tens of symbols.       

     The present invention concentrates on the areas of single carrier wireless systems with frequency domain equalizer and provides means to eliminate the inter-frame interference due to multi-path fading, and simultaneously provides coarse frame timing, carrier synchronization and channel estimation without additional overhead. 
     STATE OF THE ART 
     The frame structure of orthogonal frequency division multiplex (OFDM) systems or conventional single carrier systems with frequency domain equalizer can be seen in  FIG. 1 . The main advantage of OFDM systems, also named as multi-carrier wireless communication systems, is the low complexity frequency domain equalization. In  FIG. 2  an example of a block diagram of an OFDM systems is shown. 
     The conventional single carrier wireless system with frequency domain equalizer uses cyclic prefix for carrier synchronization. Normally the coarse frame timing and channel estimation are realized by introducing the additional pilot frame and the frame adopts constant amplitude zero auto-correlation sequence (CAZAC). 
     The disadvantages of the state of the art technology for single carrier wireless systems using frequency domain equalizer are as follows:
         Additional pilot frame overhead is required for coarse frame timing and channel estimation   Carrier synchronization using cyclic prefix is sensitive to channel impulse response       

     SUMMARY OF THE INVENTION 
     The present invention relates to a method for generating single carrier wireless communication signal, whereby said communication signal is based on a temporal frame structure, said frame structure comprising a guard interval and a data frame, said method comprising a step of inserting a cyclic prefix into said guard interval, said cyclic prefix comprising at least one pseudorandom-noise sequence. 
     Favourably, at least two of said pseudorandom-noise sequences are equal to each other. 
     Favourably, at least two of said pseudorandom-noise sequences are different to each other. 
     Favourably, said plurality of said pseudorandom-noise sequence is arranged symmetrically within the cyclic prefix. 
     Favourably, at least two of said pseudorandom-noise sequences are arranged alternatingly within the cyclic prefix. 
     Favourably, said at least one pseudorandom-noise sequence are consecutively arranged within the cyclic prefix. 
     Favourably, said cyclic prefix completely fills the guard interval. Favourably, said cyclic prefix is a part of the guard interval. 
     Favourably, a remaining part of the guard interval is situated before and/or after said cyclic prefix. 
     Favourably, said remaining part of the guard interval comprises a sequence of zeros. Favourably, at least one of said pseudorandom-noise sequences corresponds to a maximum length sequence. 
     The present invention also relates to signal generator operable to generate single carrier wireless communication signal, whereby said communication signal is based on a temporal frame structure, said frame structure being operable to provide data management and comprising a guard interval and a data frame, said transmitter comprising a cyclic prefix insertion device operable to insert a cyclic prefix into said guard interval, said cyclic prefix comprising at least one pseudorandom-noise sequence. 
     Favourably, at least two of said pseudorandom-noise sequences are equal to each other. 
     Favourably, at least two of said pseudorandom-noise sequences are different to each other. 
     Favourably, said plurality of said pseudorandom-noise sequence is arranged symmetrically within the cyclic prefix. 
     Favourably, at least two of said pseudorandom-noise sequences are arranged alternatingly within the cyclic prefix. 
     Favourably, said at least one pseudorandom-noise sequence are consecutively arranged within the cyclic prefix. 
     Favourably, said cyclic prefix completely fills the guard interval. 
     Favourably, said cyclic prefix is a part of the guard interval. 
     Favourably, a remaining part of the guard interval is situated before and/or after said cyclic prefix. 
     Favourably, said remaining part of the guard interval comprises a sequence of zeros. 
     Favourably, at least one of said pseudorandom-noise sequences is a maximum length sequence. 
     The present invention also relates to a method for processing a received single carrier wireless communication signal, whereby said communication signal is based on a temporal frame structure, said frame structure being operable to provide data management and comprising a guard interval and a data frame, whereby said guard interval comprises a cyclic prefix, said cyclic prefix comprising at least one pseudorandom-noise sequence, said method comprising the steps of correlating at least a part of said at least one pseudorandom-noise sequence of the cyclic prefix with at least one predetermined pseudorandom-noise sequence and outputting a correlation function. 
     Favourably, said method realizes coarse timing synchronization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said coarse timing synchronization of said single carrier wireless communication signal is based on the autocorrelation peak of said correlation function. 
     Favourably, said method realizes channel estimation of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said method realizes carrier synchronization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said carrier synchronization of said single carrier wireless communication signal is based on the spanned angle of two in-phase/quadrature constellation points of autocorrelation peaks of two consecutive cyclic prefixes. 
     Favourably, said carrier synchronization of said single carrier wireless communication signal is based on the phase difference rotation between said two constellation points and on the time interval between the autocorrelation peaks of said two pseudorandom-noise sequences of two consecutive cyclic prefixes. 
     Favourably, said method realizes signal-noise-ratio estimation of said single carrier wireless communication signal is based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said signal-noise-ratio estimation of said single carrier wireless communication signal is based on the autocorrelation side-lobe of said correlation function, in case the correlation function comprises an auto-correlation side-lobe. 
     Favourably, said method realizes minimum mean-square error channel equalization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said method comprises steps of applying Discrete Fourier Transformation to a channel transfer function in the time domain of said communication signal and/or of said correlation function and outputting a channel transfer function in the frequency domain, estimating signal-noise-ratio of said channel transfer function and/or of said correlation function, and applying Fast Fourier Transformation to said data frame. 
     Favourably, said method comprises steps of applying Fast Fourier Transformation to a channel transfer function in the time domain of said communication signal and/or of said correlation function and outputting a channel transfer function in the frequency domain, estimating signal-noise-ratio of said channel transfer function and/or of said correlation function, and applying Fast Fourier Transformation to said data frame. 
     Favourably, said method comprises a step of realising minimum mean-square error (MMSE) channel equalization by processing said channel transfer function in the frequency domain, said signal-noise-ratio and said Fast Fourier Transformation of said data frame. 
     The present invention also relates to a signal processor operable to process a received single carrier wireless communication signal, whereby said communication signal is based on a temporal frame structure, said frame structure being operable to provide data management and comprising a guard interval and a data frame, said guard interval comprising a cyclic prefix, said cyclic prefix comprising at least one pseudorandom-noise sequence said receiver comprising a correlation device operable to correlate at least a part of said at least one pseudorandom-noise sequence of the cyclic prefix with at least one predetermined pseudorandom-noise sequence and to output a correlation function. 
     Favourably, said signal processor is operable to realize coarse timing synchronization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said coarse timing synchronization of said single carrier wireless communication signal is based on the autocorrelation peak of said correlation function. 
     Favourably, said signal processor is operable to realize channel estimation of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or correlation function. 
     Favourably, said signal processor is operable to realize carrier synchronization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said carrier synchronization of said single carrier wireless communication signal is based on the spanned angle of two in-phase/quadrature constellation points of autocorrelation peaks of two pseudorandom-noise sequences of two consecutive cyclic prefixes. 
     Favourably, said carrier synchronization of said single carrier wireless communication signal is based on the phase difference rotation between said two constellation points and on the time interval between the autocorrelation peaks of said two consecutive cyclic prefixes. 
     Favourably, said signal processor is operable to realize signal-noise-ratio estimation of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said signal-noise-ratio estimation of said single carrier wireless communication signal is based on the auto-correlation side-lobe of said correlation function, in case the correlation function comprises an auto-correlation side-lobe. 
     Favourably, said signal processor is operable to realize minimum mean-square error (MMSE) channel equalization of said single carrier wireless communication signal based on said at least a part of said at least one pseudorandom noise sequence and/or on said correlation function. 
     Favourably, said signal processor is operable to apply Discrete Fourier Transformation to a channel transfer function in the time domain of said communication signal and/or of said correlation function and to output a channel transfer function in the frequency domain, to estimate signal-noise-ratio of said channel transfer function and/or of said correlation function, and to apply Fast Fourier Transformation to said data frame. 
     Favourably, said signal processor is operable to apply Fast Fourier Transformation to a channel transfer function in the time domain of said communication signal and/or of said correlation function and to output a channel transfer function in the frequency domain, to estimate signal-noise-ratio of said channel transfer function and/or of said correlation function, and to apply Fast Fourier Transformation to said data frame. 
     Favourably, said signal processor is operable to realise minimum mean-square error (MMSE) channel equalization by processing said channel transfer function in the frequency domain, said signal-noise-ratio and said Fast Fourier Transformation of said data frame. 
    
    
     
       DESCRIPTION OF THE DRAWINGS 
       The features, objects and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings, wherein: 
         FIG. 1  shows an example of a frame structure of OFDM systems or single carrier systems using frequency domain equalizer, 
         FIG. 2  shows an example of a block diagram of OFDM systems, 
         FIG. 3  shows an example of a block diagram of single carrier systems using frequency domain equalizer, 
         FIG. 4  shows an example of a frame structure as an embodiment of the present invention, 
         FIG. 5  shows an example of a frame structure as an embodiment of the present invention and the coarse frame timing and the carrier synchronization based on the auto-correlation peak of PN sequence, 
         FIG. 6  shows an apparatus for channel equalization being an additional part for an alternative embodiment of the present invention based on Fast Fourier Transformation (FFT), 
         FIG. 7  shows an apparatus for channel equalization being an additional part for an alternative embodiment of the present invention based on Discrete Fourier Transformation (DFT), 
         FIG. 8  shows an example of a frame structure with additional guard interval as an alternative embodiment of the present invention, 
         FIG. 9  shows an example of a flow chart comprising a data timing recovery scheme as an alternative embodiment of the present invention, 
         FIG. 10  shows an example of a I/Q constellation rotation of the strongest auto-correlation peak from two nearby PN sequences, 
         FIG. 11  shows two examples of an auto-correlation graph of two signals, 
         FIG. 12  shows two examples of an auto-correlation graph of a M sequence and a PN sequence, and 
         FIG. 13  shows an example of a guard interval and the arrangement of the cyclic prefix as well as of the PN-sequences. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     This invention describes an application/management example of a frame structure for a single carrier system with frequency domain equalizer, whereby a M (maximum length) sequence  40   b  or a PN (pseudorandom noise) sequence is used as cyclic prefix in a guard interval  44   b . Said frame structure is shown in  FIG. 4 . 
     In this invention a M sequence can be exchanged by a PN sequence and vice-versa, when not mentioned otherwise. 
     The cyclic prefix is able to cope with time dispersive multi-path fading channel, as long as the channel impulse response is shorter than the cyclic prefix. Otherwise there might be inter-frame interference. The introduced overhead is the same as or even less compared to the state of the art like conventional single carrier systems with frequency domain equalizer or OFDM systems. 
     Beside acting as cyclic prefix to eliminate the inter-frame interference as long as the channel impulse response is shorter than cyclic prefix, a PN sequence as a cyclic prefix is used to facilitate coarse timing, channel estimation and carrier synchronization for single carrier wireless systems using frequency domain equalizer. 
     The channel estimation accuracy can be improved using a consecutive PN sequence. In the following the basics of a PN sequence and a M sequence, respectively, as well as their characteristics are explained below. 
     Generally speaking, a signal comprising a message unknown to a receiver has a random nature and is called stochastic signal. In case the signal would not have a random nature, the receiver would be capable to reconstruct the message from the already sent signal due to the deterministic nature of the signal. 
     Regarding specific definitions, a signal of deterministic character is a signal, which has a value x as a real number for every time t. A signal of stochastic character is a signal, which has a random number y for every time t, whereby said number y can be presented in a probability density function. 
     Regarding the definition of an auto-correlation function φ(τ):
         said function is an even function φ(τ)=φ(−τ)   φ(τ=0) is the quadratic mean-value and therefore represents the signal power   the maximum value of said function is at τ=0       

     An ideal auto-correlation function is defined as: 
     
       
         
           
             
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     A non-ideal auto-correlation function comprises several values more, whereby an almost ideal auto-correlation function of periodic consecutive function is presented in  FIG. 12  on the left graph and a non-ideal auto-correlation function is presented on the right graph. 
     A PN sequence is a pseudo-random noise signal, which displays some deterministic features like periodic behaviour. A periodic cycle within the sequence can recur at least once. In case that the periodic cycle is as long as the PN sequence, meaning exactly one period cycle is available, said sequence is also defined as M sequence, standing for “maximum length sequence”. 
     The PN-sequence itself is characterised as follows:
         a PN-sequence comprises binary numbers; for example high value symbols like ‘1’ and low value symbols like ‘0’   PN stands for pseudo-random noise; this means that the signal is not completely random but is determinable; eventually the signal has a periodic sequence   PN-sequences can be realized by a feed-back shift register comprising m stages   The feed-back shift register comprises at least two feedbacks from any m th  stage to the first stage, whereby one feedback is always provided from the m th  stage   PN-sequences might comprise a favourable auto-correlation function   the auto-correlation function of the PN-sequence has the same period like the respective PN-sequence itself   The number of high values equals the number of low values plus one   The low value like ‘0’ cannot appear m-times in succession, thus in case of e.g. m=4 stages no sequence of 4 consecutive ‘0’ is possible   The start sequence of said feed-back shift register never comprises a ‘0’ in every stage, whereby in this case no change of the inputted values would occur   Periodic cross-correlation function of two orthogonal PN-sequences always equals zero, which is ideal for CDMA (code division multiple access) applications       

     The M-sequence is a special case of PN-sequences and is characterised as follows:
         M-sequence has a maximum period length P of a PN-sequence P=2 m −1   M-sequence offers outstanding statistical characteristics   M-sequence offers a two-valued periodical auto-correlation function, which is favourable for CDMA       

     
       
         
           
             
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             In this case the auto-correlation function of the M-sequence shows the similar behaviour of an ideal auto-correlation function as mentioned above, since only two values are part of said function. The graph is displayed in  FIG. 12  on the left side. 
             Otherwise the periodical cross-correlation function of two M-sequences is not favourable 
           
         
       
    
     The main advantages of the present invention are that:
         a PN sequence with good auto-correlation peak and small auto-correlation side-lobe is used for cyclic prefix. Compared with conventional single carrier wireless system with frequency domain equalization, the overhead introduced by cyclic prefix does not change. Since the PN sequence will be used for coarse frame timing and channel estimation, there is no need for additional pilot frames. The total overhead can be reduced.   reliable carrier synchronization can be achieved using auto-correlation peak of the PN sequence instead of conventional cyclic prefix, which is sensitive to channel impulse response.   reliable coarse timing can be achieved using PN sequence instead of additional pilot frame.   reliable channel estimation can be achieved using the auto-correlation peak of PN sequence.   a MMSE channel equalization can be achieved to improve the performance using the auto-correlation side-lobe information of PN sequence.   the channel estimation accuracy can be further improved using the consecutive PN sequences.       

       FIG. 1  shows an example of a frame structure of OFDM systems or single carrier systems using frequency domain equalizer. 
     Also in the following the key principle of OFDM is explained. One key principle of OFDM is that since low symbol rate modulation schemes (i.e. where the symbols are relatively long compared to the channel time characteristics) suffer less from intersymbol interference caused by multipath, it is advantageous to transmit a number of low-rate streams in parallel instead of a single high-rate stream. Since the duration of each symbol is long, it is feasible to insert a guard interval between the OFDM symbols, thus eliminating the intersymbol interference. 
     The guard-interval also reduces the sensitivity to time synchronization problems. 
     Although the guard interval only contains redundant data, which means that it reduces the capacity, some OFDM-based systems, such as some of the broadcasting systems, deliberately use a long guard interval in order to allow the transmitters to be spaced farther apart in a single frequency network (SFN), and longer guard intervals allow larger SFN cell-sizes. A rule of thumb for the maximum distance between transmitters in an SFN is equal to the distance a signal travels during the guard interval, for instance, a guard interval of 200 microseconds would allow transmitters to be spaced 60 km apart. 
     This frame structure  13  of a OFDM system comprises three cyclic prefixes  10   a ,  10   b ,  10   c  and three data frames  12   a ,  12   b ,  12   c  and is shown in the time domain. The basic frame structure comprises one cyclic prefix and one data frame like  10   a  and  12   a  and can be chained successively. The cyclic prefixes  10   a ,  10   b ,  10   c  are embedded in the guard intervals  14   a ,  14   b ,  14   c , respectively. At the chronological end of the respective data frames  12   a ,  12   b ,  12   c , a respective end  11   a ,  11   b ,  11   c  is designated, said ends  11   a ,  11   b ,  11   c  being part of the respective data frames  12   a ,  12   b ,  12   c.    
     In OFDM a data frame is processed by a FFT (Fast Fourier Transformation), whereby the FFT window is as long as the data frame, said FFT window determining the time when said data is being processed by the system and/or the size of the data to be transformed by FFT step by step or at once. In an OFDM symbol the cyclic prefix  10   a  is a repeat of the end of the symbol  11   a  which is placed at the beginning of said data frame  12   a.    
       FIG. 2  shows an example of a block diagram of OFDM systems. 
     Said OFDM system comprises a transmitter  33  and a receiver  34 , whereby said transmitter  33  is operable to modulate and transmit electromagnetic waves which are orthogonal frequency division multiplexed, eventually. Said receiver  34  is operable to receive electromagnetic waves and also demodulate said waves which are orthogonal frequency division multiplexed. Said OFDM system is operable to establish a wireless connection and exchange data between its transmitter  33  and receiver  34 . 
     The transmitter  33  comprises a Quadrature amplitude modulation (QAM) modulator  20 , an Inverse Fast Fourier Transformation (FFT) module  21 , a Cyclic prefix insertion module  22 , a Radio frequency transmitter  23  and an antenna  35 . The QAM modulator  20  is connected to the Inverse FFT module  21 , the Inverse FFT module  21  is connected to the Cyclic prefix insertion module  22 , the Cyclic prefix insertion module  22  is connected to the Radio frequency transmitter  23  and the Radio frequency transmitter  23  is connected to the antenna  35 . 
     First an input signal to be modulated and transmitted is sent to the QAM modulator  20 . 
     The QAM modulator  20  is operable to modulate an input signal according to QAM. The Inverse FFT module  21  is operable to apply an inverse FFT transformation on the signal received from the QAM modulator  20 . The Cyclic prefix insertion module  22  is operable to insert cyclic prefixes like  10   a ,  10   b ,  10   c  of  FIG. 1  into the signal received from the Inverse FFT module  21 . The Radio frequency transmitter  23  is operable to convert the signal received from the Cyclic prefix insertion module  22  into a signal which is transmittable by the antenna  35 , said antenna  35  being operable to transmit electromagnetic waves carrying data based on said input data. 
     The receiver  34  comprises an antenna  36 , a Radio frequency receiver  24 , a Remove cyclic prefix module  25 , a FFT module  26 , a Channel equalizer  27 , a Channel estimation module  28  and a QAM demodulator  29 . The antenna  36  is connected to the Radio frequency receiver  24 , the Radio frequency receiver  24  is connected to the Remove cyclic prefix module  25 , the Remove cyclic prefix module  25  is connected to the FFT module  26 , the FFT module  26  is connected to both the Channel equalizer  27  and the Channel estimation module  28 , the Channel estimation module  28  is additionally connected to the Channel equalizer  27  and the Channel equalizer  27  is eventually connected to the QAM demodulator  29 . 
     Finally an output signal sent out by the QAM demodulator  29  can now be further processed. 
     The antenna  36  is operable to receive the signal sent by the antenna  35  and convert said electromagnetic signal into an electric signal. The Radio frequency receiver  24  is operable to receive the electric signal from the antenna  36  and convert said signal into a baseband signal. The Remove cyclic prefix module  25  is operable to receive the signal from the Radio frequency receiver  24  and remove the inserted cyclic prefixes like e.g.  10   a ,  10   b ,  10   c  of  FIG. 1  from said signal. The FFT module  26  is operable to transform the signal received from the Remove cyclic prefix module  25  according to a Fast Fourier Transformation. The Channel estimation module  28  is operable to receive the signal from the FFT module  26  and estimate the channel quality and other characteristics based on the channel, said channel corresponding to the wireless connection between the transmitter and the receiver. The channel quality might also describe the background and/or receiver noise. The Channel equalizer  27  is operable to receive one signal sent by the FFT module  26  and one signal sent by the Channel estimation module  28 . Then the Channel equalizer  27  compensates for the dynamic frequency response of the wireless channel. The QAM demodulator  29  is operable to demodulate the signal sent by the Channel equalizer  27  and output a demodulated output signal. 
       FIG. 3  shows an example of a block diagram of single carrier systems using frequency domain equalizer. 
     Said single carrier system being also an embodiment of the present invention comprises a transmitter  31  and a receiver  32 , whereby said transmitter  31  is operable to at least modulate and transmit electromagnetic waves which are modulated onto one single carrier, eventually. Said receiver  32  is operable to at least receive electromagnetic waves and also demodulate said waves which are modulated onto one single carrier. Said single carrier system is operable to establish a wireless connection and exchange data between its transmitter  31  and receiver  32 . 
     Except for the missing Inverse FFT module  21  the transmitter  31  corresponds to the transmitter  33  shown in  FIG. 2 . Vice-versa the receiver  32  corresponds to the receiver  34  shown in  FIG. 2  and additionally comprises an Inverse FFT module  21 , the function of said module  21  corresponding to the one described in  FIG. 2  and being connected between the Channel equalizer  27  and the QAM demodulator  29  of said receiver  32 . 
     Regarding the receiver  32 , the signal from the Remove cyclic prefix module  25  is processed in the FFT module  26  step by step based on the size of the FFT frame. The FFT frame defines the time length, whereby a part of the signal is processed all at once by the FFT module  26 . 
     In another embodiment of the present invention the cyclic prefix insertion module  22  is operable to insert M sequences or PN sequences into the frame structure, which is explained later in detail. 
     The receiver  32  and the transmitter  31  could be part of one mobile wireless device. Moreover the receiver  32  and the transmitter  31  might be integrated in a semiconductor chip and comprise additional modules operable to extend the operability of the said receiver and/or transmitter, which are not shown in the  FIG. 3  for the sake of clarity. 
     The complexity of single carrier wireless systems with frequency domain equalizer is almost the same as that of OFDM wireless systems. 
     However, when compared with OFDM systems, the main advantages of single carrier wireless systems with frequency domain equalizer can be summarized as follows
         The energy of individual symbols is transmitted over the whole available frequency spectrum. Therefore, narrow band notches within the channel transfer function have only small impact on the performance. For OFDM systems, narrow band notches would degrade the performance of transmitted symbols assigned over the relevant sub-carriers. Of course, the diversity can be regained partly by utilizing error control decoder with some performance loss.   Low peak to average ratio for the radiated signal, which makes the power amplifier (PA) from the transmitted side more efficient and cheaper, especially for the millimeter wave wireless systems.   Robust to the effect of phase noise, which makes the local oscillator (LO) simpler, especially for the millimeter wave wireless systems.   The number of analogue-digital-converter (ADC) bits for the receiver side can be reduced, which is critical for high rate communications.   The carrier frequency error between the transmitter side and receiver side can destroy the orthogonality between subcarriers and introduce the inter-subcarrier interference for OFDM systems. However, it has no effect on single carrier systems with frequency domain equalizer.   It is more suitable for the user scenario where the transmitter side would be simple or low power consumption and the receiver side would be complex or relatively high power consumption, like high definition television.       

       FIG. 4  shows an example of a frame structure as an embodiment of the present invention. 
     This frame structure  43  being an embodiment of the present invention comprises three cyclic prefixes  40   a ,  40   b ,  40   c  and three data frames  42   a ,  42   b ,  42   c  and is shown in the time domain. Eventually the cyclic prefixes are embedded in guard intervals  44   a ,  44   b ,  44   c  and completely filled in said intervals, said guard intervals  44   a ,  44   b ,  44   c  being the respective time periods before the data frame periods  42   a ,  42   b ,  42   c.    
     The cyclic prefix  40   a ,  40   b ,  40   c  comprises at least one PN sequence and is operable to provide data which is necessary for the management of the data frames during the transmission and reception of electro-magnetic waves carrying said data. The cyclic prefix of the present invention can also be part of the data frame as shown in  FIG. 1 , but is favourably just an add-on in front or behind the adjacent data frame, so that no further redundant data is transmitted. 
     In  FIG. 4 , one data frame and one cyclic prefix comprising at least one PN sequence, for example  42   a + 40   b , are processed by a FFT (Fast Fourier Transformation), whereby the FFT window is as long as the length of the data frame  42   a  plus the length of the cyclic prefix  40   b . The frame structure is different from  FIG. 1 , whereby only data frame is processed by a FFT. Since the cyclic prefix  40   a  is the same as the cyclic prefix  40   b , based on the same principle of OFDM systems, the inter-frame interference introduced by the time disperse multi-path fading channel can be eliminated when the wireless channel delay is less then the length of cyclic prefix. 
     The cyclic prefix  40   a ,  40   b ,  40   c  also helps the receiver  32  to correctly place the FFT frames and indicates the beginning of the respective data frames  42   a ,  42   b ,  42   c  being processed during a respective FFT frame if one PN sequence is used as  40   a ,  40   b ,  40   c . The content of the cyclic prefixes  40   a ,  40   b ,  40   c  could be different, similar or equal to each other. 
     The guard interval  44   a ,  44   b ,  44   c  is operable to provide guard time for propagation delay and to clearly separate the respective data frames  42   a ,  42   b ,  42   c  from each other, so the data of one data frame does not overlap with data of an adjacent data frame in case of multipath propagation during transmission. 
     The data frame  42   a ,  42   b ,  42   c  is operable to provide data and/or information of any kind which is based on or corresponds to the content of a conversation like e.g. a phone call or other data meant to be transmitted and received by another communication participant. These data might comprise for example emails, pictures and the like. The data frames  42   a ,  42   b ,  42   c  are always of the same size, whereby their data does not necessarily fill out said data frames completely. 
     The sequence or alternatively said the time flow of the frame structure starts with the first cyclic prefix  40   a , continues with adjacent first data frame  42   a , then the second cyclic prefix  40   b , the second data frame  42   b , the third cyclic prefix  40   c  and ends with the third data frame  42   c.    
     Of course, the frame structure is not limited to these three data frames and three cyclic prefixes, but can go on and form a chain of frames. 
     A FFT frame, whose operability was already explained in  FIG. 1 , might be as long as the combination of at least one data frame  42   a ,  42   b ,  42   c  and of at least one cyclic prefix  40   a ,  40   b ,  40   c . This is different to  FIG. 1 , wherein one data frame is treated as one FFT frame. 
     Alternatively the FFT frame might comprise also a part of a preceding cyclic prefix, a complete succeeding data frame and a part of a succeeding cyclic prefix, like for example  40   b ,  42   b  and  40   c . if for example, several similar PN sequences are concatenated and used as cyclic prefix. This means that the FFT frame can begin somewhere in the first cyclic prefix like  40   a , covers the complete data frame  42   a  and ends somewhere in the succeeding cyclic prefix  40   b . As a result, dynamic guard interval length can be achieved. 
     As an alternative embodiment the FFT frame and/or the time of the FFT frame itself might comprise at least one data frame and one cyclic prefix. 
     Alternatively the FFT frame might comprise one data frame and the two adjacent cyclic prefixes. 
     Furthermore, adjacent FFT frames might either be situated side by side or they might overlap with each other. They could partially or completely overlap the area of the cyclic prefix and/or the guard interval, respectively, and/or each other, respectively. When two FFT frames overlap each other, two separate FFT modules might be necessary to independently read and/or process said two FFT frames, respectively. 
     In case of side by side the border between the two FFT frames might correspond with the border of the data frame and the guard interval or the border of the data frame and the cyclic prefix. Alternatively the border is situated somewhere in the guard interval. In  FIG. 8  this embodiment is later explained in more detail. 
     Regarding the PN sequence, the cyclic prefix might comprise either a single PN sequence or a plurality of identical or different PN sequences, whereby said plurality of PN sequences is formed as a continuous string of sequences. In case of different PN sequences, said continuous string might comprise a random or a deterministic pattern based on how the PN sequences are arranged within the string. One pattern might comprise two different PN sequences which alternate within the cyclic prefix. In another example the cyclic prefix comprises a symmetric arrangement of different PN sequences. These examples are described in more detail in  FIG. 13 . Depending on the pattern specific characteristics of the cyclic prefix can be read out and matched, respectively, like e.g. location within or speed/carrier synchronization of the cyclic prefix. 
     The correlation of the cyclic prefix with a predetermined and/or controllable function comprising one or a plurality of identical or different PN sequences is performed in a receiver like  32  of  FIG. 3 , operable to receive the signal sent from the transmitter  31 . The choice regarding the predetermined function and its amount and/or arrangement of PN sequences is dependent on the characteristics of the correlation of the cyclic prefix to be determined. 
       FIG. 5  shows an example of a frame structure as an embodiment of the present invention and the coarse frame timing and the carrier synchronization based on the auto-correlation peak of PN sequence. 
     This frame structure  43  corresponds to the frame structure  43  shown in  FIG. 4  and comprises four cyclic prefixes  40   a ,  40   b ,  40   c ,  40   d  and three data frames  42   a ,  42   b ,  42   c , whereby said cyclic prefixes  40   a ,  40   b ,  40   c ,  40   d  are or comprises maximum length (M) sequences or pseudorandom noise (PN) sequences. Below each of these PN sequences  40   a ,  40   b ,  40   c  the correlation function of said PN sequences is shown as a graph  53   a ,  53   b ,  53   c , respectively. 
     The correlation graphs  53   a ,  53   b ,  53   c  of the PN sequences comprises a high correlation peak and a low auto-correlation side-lobe, respectively, as later shown in  FIG. 11 . This auto-correlation function is created in a receiver, when the signal with the frame structure comprising the PN sequence is received and correlated with an identical PN sequence. 
     In case the received cyclic prefix  40   a  comprises a plurality of identical PN sequences formed as a continuous string and is auto-correlated with one identical PN sequence at a receiver, the auto-correlation graph of the PN sequence will comprise a plurality of high correlation peaks and low auto-correlation side-lobes. 
     In another example the cyclic prefix  40   a  comprises a plurality of PN sequences formed as a continuous string and is auto-correlated with one PN sequence being part of said string, it is possible to locate the exact position within the cyclic prefix, when the high correlation peak appears in the graph. 
     Instead of one single PN sequence, a correlation sequence of identical or different PN sequences is used for correlating with said received cyclic prefix  40   a , whereby said correlation sequence or a plurality of said correlation sequence are part of said received cyclic prefix  40   a.    
     Due to the characteristics of the correlation graphs  53   a ,  53   b ,  53   c  of the PN sequences  40   a ,  40   b ,  40   c , the PN sequence is used to realize coarse timing, channel estimation carrier synchronization, obtain signal-noise-ratio (SNR) estimation and/or implement minimum mean-square error (MMSE) channel equalization. The MMSE channel equalization is described more in detail in  FIG. 6  or  7 . 
     Based on the characteristics of the graphs  53   a ,  53   b ,  53   c , it is possible to determine the beginning of the FFT frame. The FFT frame might start from the beginning or at the end of the graphs  53   a ,  53   b ,  53   c . Also the high correlation peak or the low auto-correlation side-lobe might be the starting point of the FFT frame. The FFT frame, which is already explained in  FIG. 4 , comprises at least the data frame succeeding the respective PN sequence. Alternatively the beginning of the FFT frame is independent from the cyclic prefix and/or the guard interval, but at least comprises the complete succeeding data frame. 
     In particular the coarse frame timing can be determined by the auto-correlation peak of the graph  53   a ,  53   b ,  53   c  of the PN sequence as shown in  FIG. 5 . 
     The carrier synchronization can be implemented based on I/Q constellation rotation of the strongest auto-correlation peak from two nearby PN sequence. Below the correlation graphs  53   a  and  53   b  the respective constellation points  51  and  52  are shown in a Cartesian coordinates. The phase difference between these two constellation points and the time period between the two PN sequences  40   a  and  40   b  can be used for carrier synchronization. The I/Q constellation rotation is shown in detail in  FIG. 10 . 
     The cyclic prefix might comprise at least one pseudorandom-noise sequence, whereby said one pseudorandom-noise sequence is complex value and comprises one I-channel pseudorandom-noise sequence and one Q-channel pseudorandom-noise sequence. In alternative embodiments the I-channel sequence and the Q-channel sequence could either be the same or different to each other. 
     The channel transfer function can be estimated based on several auto-correlation peaks of the graph  53   a ,  53   b ,  53   c  of the PN sequence, whereby the auto-correlation side-lobe from PN sequence can be used for signal to noise ratio (SNR) calculation. The acquired information can be used for MMSE channel equalization. 
       FIG. 6  shows an apparatus for channel equalization being an additional part for an alternative embodiment of the present invention based on Fast Fourier Transformation (FFT). 
     This apparatus comprises a FFT module  65 , a SNR estimation module  62 , a FFT module  63  and a MMSE channel equalization module  64 , whereby said apparatus is operable for channel equalization. The channel equalization is mainly used in a receiver like  32  of  FIG. 3 . 
     At least a part of said apparatus can be implemented into the receiver  32  of  FIG. 3  as channel equalizer  27 ; in particular the MMSE channel equalizer  64  can be implemented as said equalizer  27 . 
     The FFT module  65  is operable to receive a signal which is a channel transfer function in the time domain, convert said signal into a channel transfer function in the frequency domain and output said signal. The SNR estimation module  62  is operable to receive the same channel transfer function in the time domain, which was received by the FFT module  65  and calculate and/or estimate the signal-noise-ratio of said function. The FFT module  63  is operable to receive a signal comprising the data frame and apply the FFT to said signal. The MMSE channel equalization module  64  is operable to receive the channel transfer function in the frequency domain provided by the FFT module  65 , the SNR estimation signal provided by the SNR estimation module  62  and the signal provided by the FFT module  63  and eventually calculate and demodulate the output signal. 
     It has to be ensured that the channel transfer function  53  comprises the PN sequence with a main high auto-correlation lobe and a smaller auto-correlation side-lobe. 
       FIG. 7  shows an apparatus for channel equalization being an additional part for an alternative embodiment of the present invention based on Discrete Fourier Transformation (DFT). 
       FIG. 7  shows an apparatus for channel equalization being an additional part for an alternative embodiment of the present invention based on Discrete Fourier Transformation (DFT). 
     This apparatus comprises a Discrete Fourier Transformation (DFT) module  61 , a SNR estimation module  62 , a FFT module  63  and a MMSE channel equalization module  64 , whereby said apparatus is operable for channel equalization. 
     Except for the missing FFT module  65  the apparatus of  FIG. 7  corresponds to the apparatus of  FIG. 6 . Both apparatuses can be implemented into the receiver. 
     Like in  FIG. 6 , it has to be ensured in  FIG. 7  that the channel transfer function  53  comprises the PN sequence with a main high auto-correlation lobe and a smaller auto-correlation side-lobe. 
     As shown in  FIG. 6 , FFT can be used instead of DFT to reduce the calculation complexity for obtaining the channel transfer function from frequency domain, which will be adopted for channel equalization. 
       FIG. 8  shows an example of a frame structure with additional guard interval as an alternative embodiment of the present invention. 
     This frame structure is based on the frame structure  43  shown in  FIG. 4  and comprises three cyclic prefixes  80   a ,  80   b ,  80   c  and three data frames  82   a ,  82   b ,  82   c , whereby said cyclic prefixes  80   a ,  80   b ,  80   c  are or comprise maximum length (M) sequences or pseudorandom noise (PN) sequences. Between each data frame  82   a ,  82   b ,  82   c  a respective guard interval  83   a ,  83   b ,  83   c  exists. In each of said guard intervals  83   a ,  83   b ,  83   c  a respective PN sequence  80   a ,  80   b ,  80   c  is embedded. Since the guard intervals  83   a ,  83   b ,  83   c  are in this embodiment larger than the PN sequences  80   a ,  80   b ,  80   c , some free space is left on the right and left side of the PN sequences  80   a ,  80   b ,  80   c . For example and in detail the first free space  84   a  is located between the data frame  82   a  and the PN sequence  80   b  and the second free space  84   b  is located between the PN sequence  80   b  and the data frame  82   b.    
     Thus, the guard interval  83   a ,  83   b ,  83   c  between the PN sequence and the data frame can be extended. If the length of guard interval  83   a ,  83   b ,  83   c  is longer than the wireless channel delay spread, there is no effect on the correlation peak from the data frame part and more accurate channel estimation can be obtained. 
     The further guard interval between the PN sequence and the data frame, meaning the first and/or second free space  84   a  and  84   b  can comprise a sequence of zeros. The two free spaces  84   a  and  84   b  might be of different or equal size, respectively. 
       FIG. 9  shows an example of a flow chart comprising a data timing recovery scheme as an alternative embodiment of the present invention. 
     In detail the flow chart is a timing offset compensation scheme and comprises nines steps S 1  to S 9 , which is another way of the frequency-domain equalization with cyclic prefixes. 
     The FFT frame comprises the data frame and the cyclic prefix as explained in  FIG. 4  and is read out in step S 1 . In the next step S 2  a FFT is applied to the signal from S 1 , said signal comprising the FFT frame. In S 3  a frequency-domain equalization is conducted to the signal received from step S 2 . In S 4  an Inverse FFT is applied on the signal from S 3 . 
     In step S 5  a decision is made based on the preceding steps whereby the (hard or soft) decision data is output. After the decision step S 5  the data comprising the data frame and a part of the preceding and succeeding cyclic prefix is the result of step S 5  and is finally determined and shown in step S 6 , respectively. After making the decision data output loop in step S 7 , the correlator searches and eventually determines the border of cyclic prefix and data by applying a PN sequence with said decision output for correlation in step S 8  and finally the data part can be derived and shown in step S 9 . 
     According to the invention, the timing of FFT frame, meaning the beginning of said FFT frame, does not need to be on the exact place for every FFT frame, but can be placed individually within or on the borders of the cyclic prefix or guard interval as explained above. 
     In case of high rate mm-wave system, the timing offset change over the data frame becomes comparable with the data symbol length, because of the absolute value of clock offset. By having such idea, the timing offset can be adjusted finally. 
       FIG. 10  shows an example of a I/Q constellation rotation of the strongest auto-correlation peak from two nearby PN sequences. 
     There are two constellation points  51   a  and  52   a  of two respective PN sequences shown in a complex plane of a coordinate system, also called constellation diagram. From the view of the origin of the coordinate system the two constellation points  51   a  and  52   a  envelop an angle α. 
     As the symbols are represented as complex numbers, they can be visualized as points on the complex plane. The real and imaginary axes are often called the inphase, or I-axis and the quadrature, or Q-axis. Plotting several symbols in a scatter diagram produces the constellation diagram. The points on a constellation diagram are called constellation points. 
     The constellation points  51   a  and  52   a  are based on the constellation points  51  and  52  shown in  FIG. 5 . 
       FIG. 11  shows two examples of an auto-correlation graph of two signals. 
     The left graph as well as the right graph are auto-correlation graphs, respectively, and are symmetric along the respective axis of the ordinate. Both graphs show the highest peak value at t=0, at the axis of the ordinate. The graphs are shown in a grid system on the x-axis from −1,5 to 1,5 and on the y-axis from −0,4 to 1,0, whereby the vertical lines are spaced apart by 0,25 and the horizontal lines are spaced apart by 0,2. This values are normalized values and are not restricted to these values. 
     The two signals creating the respective autocorrelation graphs can be PN sequences, whereby the resolution regarding the number of stages of the feed-back shift register is very high. 
       FIG. 12  shows two examples of an auto-correlation graph of a M sequence and a PN sequence 
     The respective grid systems are based on the grid systems shown in  FIG. 11 , except for the fact that the vertical lines are spaced apart by 0,5 and the horizontal lines are spaced apart by 0,2. Also both graphs show a symmetry at the axis of the ordinate. 
     The left graph is the auto-correlation function of a M sequence. This correlation graph displays a periodic function which comprises a high peak value  54   c  and a lower value  57 , and is thus similar to an ideal auto-correlation function of a random signal having only two values. Since the left graph comprises a plurality of identical M sequences, a plurality of identical peaks being spaced apart by a period length  58  of 1 are visible. In the case of M sequences the periodic intervals of the autocorrelation peaks equals the period length of the M sequence. 
     The right graph is the auto-correlation function of a PN sequence. This correlation graph displays a plurality of periodic peaks of value 1 and a plurality of values around y=−0,2 and 0 between the peaks. Although high peak values  54   d  are shown and are spaced apart from each other by a periodic interval  59 , the values during said periodic interval  59  are not constant like in the left graph during the interval  58 . 
       FIG. 13  shows an example of a guard interval and the arrangement of the cyclic prefix as well as of the PN-sequences. 
     The guard interval  94  comprises a boarder  95   a  between the preceding data frame and itself as well as a boarder  95   b  between the succeeding data frame and itself. Moreover the guard interval comprises a cyclic prefix  90  and two free spaces  91   a  and  91   b . The first free space  91   a  is placed between the first boarder  95   a  and the cyclic prefix  90  and the second free space  91   b  is placed between the second boarder  95   b  and the cyclic prefix  90 . The cyclic prefix comprises a symmetric axis  93  and eight PN-sequences  1  to  8  which are arranged consecutively. The unnumbered spaces at the edge and within the cyclic prefix  90  might comprise again a PN-sequence of the same length like e.g. PN-sequence  1 . 
     All components of  FIG. 13  correspond to the components described above and already shown in the other Figures. 
       FIG. 13  is basically describing the arrangement of said components. 
     In one example the cyclic prefix  90  comprises two different PN-sequences, which are arranged alternatingly, like e.g. one sequence is placed at the even positions  2 ,  4 ,  6 ,  8  and the other sequence is placed at the odd positions  1 ,  3 ,  5 ,  7 . 
     In another example the cyclic prefix comprises different PN-sequences, which are arranged symmetrical. This means e.g. that one sequence is placed at position  4  and  5 , the second sequence is placed at position  3  and  6 , the next sequence is placed at position  2  and  7 , and so on. 
     In another example the PN-sequences are not limited to be arranged consecutively but could comprise spaces between each other. These spaces might be filled with a sequence of zeros. 
     Another possiblity in view of the arrangement is considering the position of the cyclic prefix  90  with the guard interval  94 . 
     As shown the guard interval  94  comprises two free spaces  91   a  and  91   b  before and after the cyclic prefix. 
     In another example the cyclic prefix  90  extends till the first boarder  95   a  and/or the second boarder  95   b , so that only one or no free space exists. 
     In another embodiment at least two of said pseudorandom-noise sequences are equal to each other or at least two of said pseudorandom-noise sequences are different to each other. 
     Favourably, at least two pseudorandom-noise sequences are arranged symmetrically within the cyclic prefix. 
     The invention is not limited to the embodiment shown and described above by way of example, but can instead undergo modifications within the scope of the patent claims attached and the inventive concept. 
     Further embodiments of the invention are possible, but not shown in the drawings for the sake of clarity. 
     Reference Numbers 
     
       
         
           
               
               
               
             
               
                   
                   
               
             
            
               
                   
                 1 
                 First position in cyclic prefix 
               
               
                   
                 2 
                 Second position in cyclic prefix 
               
               
                   
                 3 
                 Third position in cyclic prefix 
               
               
                   
                 4 
                 Fourth position in cyclic prefix 
               
               
                   
                 5 
                 Fifth position in cyclic prefix 
               
               
                   
                 6 
                 Sixth position in cyclic prefix 
               
               
                   
                 7 
                 Seventh position in cyclic prefix 
               
               
                   
                 8 
                 Eight position in cyclic prefix 
               
               
                   
                 10a-c 
                 Cyclic prefix 
               
               
                   
                 11a-c 
                 End of data frame 1-3 
               
               
                   
                 12a-c 
                 Data frame 1-3 
               
               
                   
                 13 
                 Frame structure of state of the art 
               
               
                   
                 14a-c 
                 Guard intervals 
               
               
                   
                 20 
                 Quadrature amplitude modulation (QAM) modulator 
               
               
                   
                 21 
                 Inverse Fast Fourier Transformation (Inverse FFT) module 
               
               
                   
                 22 
                 Cyclic prefix insertion module 
               
               
                   
                 23 
                 Radio frequency transmitter (Tx RF) 
               
               
                   
                 24 
                 Radio frequency receiver (Rx RF) 
               
               
                   
                 25 
                 Remove cyclic prefix module 
               
               
                   
                 26 
                 Fast Fourier Transformation (FFT) module 
               
               
                   
                 27 
                 Channel equalizer 
               
               
                   
                 28 
                 Channel estimation module 
               
               
                   
                 29 
                 QAM demodulator 
               
               
                   
                 31 
                 Transmitter of single carrier system 
               
               
                   
                 32 
                 Receiver of single carrier system 
               
               
                   
                 33 
                 Transmitter of OFDM system (orthogonal frequency  
               
               
                   
                   
                 division multiplex) 
               
               
                   
                 34 
                 Receiver of OFDM system (orthogonal frequency  
               
               
                   
                   
                 division multiplex) 
               
               
                   
                 35 
                 Antenna of transmitter 
               
               
                   
                 36 
                 Antenna of receiver 
               
               
                   
                 40a-d 
                 PN sequence as cyclic prefix 
               
               
                   
                 42a-c 
                 Data frame 1-3 
               
               
                   
                 43 
                 Frame structure comprising an embodiment of the  
               
               
                   
                   
                 present invention 
               
               
                   
                 44a-d  
                 Guard intervals 
               
               
                   
                 51 
                 Constellation point of PN sequence of data frame 1 
               
               
                   
                 51a 
                 First constellation point of PN sequence 
               
               
                   
                 52 
                 Constellation point of PN sequence of data frame 2 
               
               
                   
                 52a 
                 Second constellation point of PN sequence 
               
               
                   
                 53 
                 Channel transfer function (time domain) 
               
               
                   
                 53a-c  
                 Correlation function of PN sequence as graph 
               
               
                   
                 54a-d 
                 High auto-correlation peak 
               
               
                   
                 55a-b 
                 Time interval between main- &amp; side-lobe 
               
               
                   
                 56a-b  
                 Low auto-correlation side-lobe 
               
               
                   
                 57 
                 Low auto-correlation value 
               
               
                   
                 58 
                 Time interval between high auto-correlation peaks 
               
               
                   
                 59 
                 Time interval between two main-lobes 
               
               
                   
                 61 
                 Discrete Fourier Transformation module 
               
               
                   
                 62 
                 Signal-Noise-Ratio estimation module 
               
               
                   
                 63 
                 Fast Fourier Transformation module 
               
               
                   
                 64 
                 Minimum mean-square error estimation module 
               
               
                   
                 65 
                 Fast Fourier Transformation module 
               
               
                   
                 80a-c  
                 PN sequence as cyclic prefix 
               
               
                   
                 82a-c  
                 Data frame 1-3 
               
               
                   
                 83a-c  
                 Guard interval 
               
               
                   
                 84a 
                 First free space 
               
               
                   
                 84b 
                 Second free space 
               
               
                   
                 90 
                 Cyclic prefix 
               
               
                   
                 91a 
                 First free space 
               
               
                   
                 91b 
                 Second free space 
               
               
                   
                 93 
                 Symmetric axis of cyclic prefix 
               
               
                   
                 94 
                 Guard interval 
               
               
                   
                 95a 
                 Boarder between preceding data frame and guard interval 
               
               
                   
                 95b 
                 Boarder between succeeding data frame and guard interval 
               
               
                   
                 S1 
                 Step of receiving data stream 
               
               
                   
                 S2 
                 Step of Fast Fourier Transformation 
               
               
                   
                 S3 
                 Step of equalization 
               
               
                   
                 S4 
                 Step of Inverse Fast Fourier Transformation 
               
               
                   
                 S5 
                 Step of decision 
               
               
                   
                 S6 
                 Decision output 
               
               
                   
                 S7 
                 Step of making the decision output loop 
               
               
                   
                 S8 
                 Step of correlating with PN sequence 
               
               
                   
                 S9 
                 Data output