Patent Publication Number: US-9419531-B2

Title: Forward-flyback DC-DC converter using resonant LC output circuit

Description:
TECHNICAL FIELD 
     The present invention relates to the field of switching power supply, more particularly, to a direct current-direct current (DC-DC) converter. 
     BACKGROUND 
     The DC-DC converter has been commercialized and widely used in UPS systems, battery charge and discharge devices, electric vehicles, starter/generator systems, aeronautics and space power systems, remote and data communication systems, computer equipment, office automation equipment, industrial instruments and meters, and other occasions. Varying with the operating mode, DC-DC converters can be divided into topological structures such as step-down, step-up, step-down/step-up, flyback, forward, half-bridge, full-bridge, push-pull, etc. With increasing requirement for the switching power supply performance, it is necessary to develop a new circuit topological structure to implement a high-efficiency DC-DC converter. 
     Due to the advantages such as low cost and wide range of input voltage, an active-clamped flyback DC-DC converter is usually applied in the “Super Charger” included in the UPS system for charging the external battery. The topological structure of the active-clamped flyback DC-DC converter in the prior art is for example shown in  FIG. 1 . The disadvantage of such converter topology rests with difficulty to meet the requirement of high efficiency (for example, an efficiency of above 94%). 
     For the need to improve the efficiency of the converter, Patent Document 1 (CN 101692595 A) proposes a forward-flyback DC-DC converter topological structure, which is shown in the schematic diagram of  FIG. 2 . 
     In order to implement the high-efficiency DC-DC converter at low cost, there is still room for further improving the circuit topological structure. 
     SUMMARY 
     The present invention is developed to solve the problem mentioned above. With the forward-flyback DC-DC converter topology of the present invention, an even higher efficiency can be implemented at the cost approximate to that of the topology in Patent Document 1. 
     According to one embodiment of the present invention, a forward-flyback DC-DC converter topology includes a transformer, a main switch, a clamp circuit, first and second rectifying switches, an LC resonant circuit and an output capacitor. A primary winding of the transformer and the main switch are connected in series between a first input terminal and a second input terminal. The clamp circuit constituted by a clamp capacitor and a clamp switch connected in series is connected in parallel with the primary winding or the main switch. A secondary winding of the transformer includes a forward winding and a flyback winding. A terminal of the primary winding through which current flows into is a dotted terminal of the primary winding, and a connecting mode of a secondary side of the transformer is: the dotted terminal of the forward winding being connected with a first output terminal via the first rectifying switch, a dotted terminal of the flyback winding being connected with a second output terminal via the second rectifying switch, the LC resonant circuit being connected with the first output terminal, the second output terminal and an unlike terminal of the forward winding and the flyback winding so that the first and second rectifying switches implement zero-current switching, and the output capacitor being connected between the first output terminal and the second output terminal. 
     Preferably, the LC resonant circuit includes a first capacitor, a second capacitor and a resonant inductor. The first capacitor and the second capacitor are connected in series between the first output terminal and the second output terminal, one terminal of the resonant inductor is connected to the unlike terminal of the forward winding and the flyback winding, and the other terminal thereof is connected to an intermediate node of the first capacitor and the second capacitor 
     Preferably, the LC resonant circuit includes a first inductor, a second inductor, the first capacitor and the second capacitor. The first inductor and the first capacitor are connected in series between the first output terminal and the unlike terminal of the forward winding and the flyback winding, and the second inductor and the second capacitor are connected in series between the second output terminal and the unlike terminal of the forward winding and the flyback winding. 
     Preferably, the LC resonant circuit includes the first inductor, the second inductor, the first capacitor and the second capacitor. The first inductor is connected between the first rectifying switch and the first output terminal, the second inductor is connected between the second rectifying switch and the second output terminal, the first capacitor and the second capacitor are connected in series between the first output terminal and the second output terminal, and the unlike terminal of the forward winding and the flyback winding is connected to the intermediate node of the first capacitor and the second capacitor. 
     Preferably, a turns ratio between the forward winding and the flyback winding is 1:1. 
     Preferably, on condition that a DC-DC power transmission of the converter in a forward working state is greater than a DC-DC power transmission in a flyback working state, the number of turns of the flyback winding is made greater than the number of turns of the forward winding. On the contrary, on condition that the DC-DC power transmission of the converter in the flyback working state is greater than the DC-DC power transmission in the forward working state, the number of turns of the forward winding is made greater than the number of turns of the flyback winding. 
     Preferably, the rectifying switch is a diode or a MOSFET. 
     Preferably, the transformer has a leakage inductance. 
     Preferably, when entering the forward working state with the main switch on and the clamp switch off, the first rectifying switch is on, the second rectifying switch is off, the LC resonant circuit begins to resonate; before the main switch is switched off, resonance current flowing through the LC resonant circuit is made to zero so as to implement the zero-current switching of the first rectifying switch. When entering the flyback working state with the main switch off and the clamp switch on, the first rectifying switch is off, the second rectifying switch is on, the LC resonant circuit begins to resonate; before the main switch is switched on, the resonance current flowing through the LC resonant circuit is made to zero so as to implement the zero-current switching of the second rectifying switch. 
     Preferably, when the main switch is off, the clamp capacitor and the leakage inductance of the transformer begin to resonate, so that the main switch and the clamp switch obtain zero-voltage switching, energy of the leakage inductance of the transformer is transferred to the secondary side via resonance, so as to avoid energy loss of the leakage inductance of the transformer and instantly-caused voltage spike on the main switch. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings which are incorporated in and constitute part of this description, show the embodiments of the present invention, and with the above-given general description and the following detailed description of the embodiments of the present invention, are used to illustrate the principle of the present invention. In the accompanying drawings: 
         FIG. 1  shows an equivalent circuit diagram of the active-clamped flyback DC-DC converter topology according to the prior art; 
         FIG. 2  shows an equivalent circuit diagram of the active-clamped forward-flyback DC-DC converter topology according to the prior art; 
         FIG. 3  shows an equivalent circuit diagram of the active-clamped forward-flyback DC-DC converter topology according to one embodiment of the present invention; 
         FIGS. 4( a )-( d )  show a set of exemplary active-clamped forward-flyback DC-DC converter topological structures according to the embodiment of the present invention; 
         FIG. 5  shows operating waveforms of the active-clamped forward-flyback DC-DC converter topological structures according to the embodiment of the present invention; 
         FIG. 6  is a curve diagram showing the efficiency comparison among the three topologies in  FIGS. 1, 2 and 3  in the case of change of the output current; 
         FIG. 7  is a curve diagram showing the efficiency comparison between the two topologies in  FIG. 2  and  FIG. 3  in the case of load variation; 
         FIG. 8  shows a first variant of the secondary-side resonant circuit; 
         FIG. 9  shows a second variant of the secondary-side resonant circuit. 
     
    
    
     DETAILED DESCRIPTION 
     The preferable embodiments according to the present invention are described below with reference to the accompanying drawings, in which like reference signs indicate like components, and therefore detailed description thereof will not be repeated, wherein “U” and “V” are both signs representing voltage, which are not used distinctively hereinafter. 
       FIG. 3  shows an equivalent circuit diagram of the active-clamped forward-flyback DC-DC converter topology according to one embodiment of the present invention. In  FIG. 3 , Lr and Lm represent leakage inductance and magnetizing inductance separated from a practical transformer equivalent model, respectively, the transformer being an ideal transformer. It can be seen from  FIG. 3  that the DC-DC converter topology according to the embodiment includes a high-frequency transformer, a main switch transistor T 1 , an active clamp circuit, rectifying diodes D 1  and D 2 , a resonant circuit and an output capacitor C 0 . 
     It can be seen from  FIG. 3  that the primary-side structure of the transformer including the active clamp circuit is the same as that in the prior art, i.e., the primary winding of the transformer (represented by Np in  FIG. 3 ) and the main switch transistor T 1  are connected in series between a first input terminal and a second input terminal. The clamp circuit constituted by a clamp capacitor Cr and a clamp switch transistor T 2  connected in series is connected to the primary winding Np in parallel. The clamp capacitor Cr resonates with a leakage inductance Lr when the main switch transistor T 1  is off, so that the main switch transistor T 1  and the clamp switch transistor T 2  obtain zero-voltage switching, and the energy of the leakage inductance Lr is transferred to the secondary side via resonance, so as to avoid energy loss of the leakage inductance Lr and instantly-caused voltage spike on the main switch transistor T 1 . 
     Alternatively, the clamp circuit constituted by the clamp capacitor Cr and the clamp switch transistor T 2  connected in series may be connected with the main switch transistor T 1  in parallel, other than connected with the primary winding Np in parallel. 
     It can be seen from  FIG. 3  that in addition to the rectifying circuit and a filter circuit, the secondary side of the transformer also includes a resonant circuit, which is constituted by a resonant inductor Ls, a first capacitor C 1  and a second capacitor C 2 , for implementing zero-current switching of the rectifying diodes D 1  and D 2 . As shown in  FIG. 3 , the secondary winding of the transformer includes a winding where current flows in the forward working state (briefly referred to as “forward winding”, represented by Ns 1  in  FIG. 3 ) and a winding where current flows in the flyback working state (briefly referred to as “flyback winding”, represented by Ns 2  in  FIG. 3 ). A terminal of the primary winding Np through which current flows into is a dotted terminal of the primary winding Np, thus, a connecting mode of the secondary side of the transformer is: the dotted terminal of the forward winding Ns 1  being connected with a first output terminal via the first rectifying diode D 1 , the dotted terminal of the flyback winding Ns 2  being connected with a second output terminal via the second rectifying diode D 2 , the first capacitor C 1  and the second capacitor C 2  being connected in series between the first output terminal and the second output terminal, one terminal of the resonant inductor Ls being connected with an unlike terminal of the forward winding Ns 1  and the flyback winding Ns 2 , and the other terminal thereof being connected to an intermediate node of the first capacitor C 1  and the second capacitor C 2 , the output capacitor C 0  being connected between the first output terminal and the second output terminal. 
     Although the secondary-side rectifying switch as shown in  FIG. 3  is a diode, those skilled in the art may conceive that a MOSFET or like switching element may be used as the secondary-side rectifying switch and the switch timing thereof may be appropriately controlled.  FIGS. 4( a )-( d )  show a set of exemplary active-clamped forward-flyback DC-DC converter topological structures according to the embodiment of the present invention, wherein  FIGS. 4( c ) and 4( d )  show the case where the clamp circuit and the main switch transistor T 1  are connected in parallel, and  FIGS. 4( b ) and 4( d )  show the case where the MOSFET is used as a rectifying switch. 
       FIG. 5  shows signal waveforms of the forward-flyback DC-DC converter in operation according to the embodiment of the present invention, wherein, S T1  and S T2  respectively represent trigger signals of the main switch transistor T 1  and the clamp switch transistor T 2 , i m  represents a waveform of the magnetizing current, i 1  represents a primary current waveform, i s  represents a resonant current flowing through the resonant inductor Ls, U T1  and i T1  respectively represent a voltage and a current waveform of the main switch transistor T 1 , U T2  and i T2  respectively represent a voltage and a current waveform of the clamp switch transistor T 1 , U D1  and -U D2  respectively represent voltage waveforms of the first and the second rectifying diodes D 1  and D 2 . When the main switch transistor T 1  is on and the clamp switch transistor T 2  is off (in forward working state), on the secondary side, the first rectifying diode D 1  is on, the second rectifying diode D 2  is off, the resonant circuit constituted by the first capacitor C 1 , the second capacitor C 2  and the resonant inductor Ls begins to resonate, half of the resonant current flows through the output capacitor C 0 , to supply power to the load connected between the first and the second output terminals. Before the main switch transistor T 1  is switched off, the resonant period Tr (=2π√{square root over (L S (C 1 +C 2 ))}) is completed by half, the resonant current i s  flowing through the resonant inductor Ls turns to be zero, thus, the first rectifying diode D 1  is switched off without reverse recovery. When the main switch transistor T 1  is off and the clamp switch transistor T 2  is on (in flyback working state), the zero-current switching of the second rectifying diode D 2  is implemented in the same way. 
     For the first and the second rectifying diodes D 1  and D 2 , the reverse voltage V RD1  of the first rectifying transistor in the flyback working state and the reverse voltage V RD2  of the second rectifying diode in the forward working state are respectively: 
     
       
         
           
             
               
                 
                   
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     Where V 0  is an output voltage, V S1  and V S2  are respectively voltages of the forward winding and the flyback winding, V in  is an input voltage, Vr is a voltage of the clamp capacitor, N P  is the number of turns of the primary winding of the transformer, N S1  and N S2  are respectively the number of turns of the forward winding and the flyback winding, and Lr may be omitted since Lm is far greater than Lr. 
     It can be seen in formulae (1) and (2) that, if N S1 =N S2 , then V RD1 =V RD2 =V 0 . In contrast, in the flyback DC-DC converter topology as shown in  FIG. 1 , the reverse voltage on the secondary-side rectifying diode is V 0 +Vin/n, where, n is the turns ratio between the primary winding and the secondary winding. It can be seen by comparison that in the forward-flyback DC-DC converter topology according to the embodiment of the present invention, a rectifying diode of lower voltage rating may be selected. 
     It may be conceived that further optimization can be made to the converter topology by using the relation between the reverse voltage of the secondary-side rectifying diode and the number of turns of the forward and flyback windings. In a more advantageous embodiment, the converter can be made to transfer different amounts of energy in the forward working state and the flyback working state, according to different secondary-side output voltage. For example, if most of the energy is transferred in the forward working state, since the current flowing through the first rectifying diode D 1  will be much greater than the current flowing through the second rectifying diode D 2 , by setting the number of turns of the secondary winding as N S2 &gt;N S1 , the reverse voltage of the first rectifying diode D 1  may be further reduced to be less than V 0 , so as to further reduce the secondary-side loss by using the diode with smaller reverse withstand voltage (the forward on-state voltage drop is relatively small accordingly) as the first rectifying diode D 1 . In other words, by adjusting the size relation between the number of turns of the secondary-side forward winding and the number of turns of the flyback winding, the reverse voltage of the secondary-side rectifying diode may be made different, so as to select the most preferable rectifying tube having the most suitable reverse withstand voltage performance, and optimize the efficiency. 
     It is easy for those skilled in the art to design and select parameters of the high-frequency transformers, inductors, capacitors, and semiconductor switching devices on the basis of the topological structure as shown in  FIG. 3 , and to design and select a control module, a drive module, a sampling circuit, and other peripheral circuits to produce a forward-flyback DC-DC converter. For example, the control module of the DC-DC converter may utilize a PWM modulation dedicated chip, into which an oscillator, an error comparator, a PWM modulator, a driving circuit and/or protection circuit are integrated. A stable and simply-controlled switching power supply may be constituted just by an integrated chip plus a few circuits. Since the design of the control chip and the peripheral components belongs to the common knowledge of those skilled in the art, detailed description thereof will be omitted here. 
     Those skilled in the art can clearly understand that, by using the forward-flyback DC-DC converter topology according to the embodiment of the present invention, the zero-current switching of the secondary-side rectifying diodes D 1  and D 2  is implemented and both the secondary-side loss and electromagnetic interference emission are reduced via the secondary-side resonant circuit; meanwhile, because the forward-flyback topology has a lower peak current on the primary side, the conduction loss of the primary-side semiconductor device may be reduced thereby. With the above factors taken together, the forward-flyback DC-DC converter topology according to the embodiments of the present invention has a higher efficiency than that of a flyback topology, and can meet the requirement for the fan-off application scenario. Meanwhile, because the reverse voltage of the secondary-side rectifying diode is relatively low, a device of lower rating may be selected so as to reduce cost. Efficiency comparison between the two topologies under test conditions that the input voltage Vin=360V DC, the output power P 0 =480 W, the output voltage V 0 =40, 60, 80, 96, 120, 160, 240, 320V DC is shown in  FIG. 6 . It can be seen in  FIG. 6  that the efficiency can be improved by up to 3%. In addition, it may be noted that, the lower the output current (the higher the output voltage), the higher the efficiency of the forward-flyback DC-DC converter topology according to the embodiment of the present invention. Therefore, the topology disclosed in the present invention is especially suitable for the application scenario of high output voltage. 
     Table 1 shows an instance in which a secondary-side diode is selected when P 0 =480 W. As shown in Table 1, a suitable secondary-side rectifying diode may be selected as the output voltage varies, wherein, the diode having repetitive reverse peak voltage (VRRM) parameters of 200, 400, 600 and 800V is selected for the flyback topology as shown in  FIG. 1 , the diode having reverse peak voltage (VRRM) parameters of 100, 200 and 400 is selected for the topology according to the present invention as shown in  FIG. 3 , to meet the requirement for different maximum reverse voltage (Max.Rev.Vol.). 
     In order to facilitate comparison, the cases of the forward-flyback converter topology in the prior art as shown in  FIG. 2  are also shown in  FIG. 6  and Table 1. In such topology, because most of the current flows through D 1  and D 4  and the current in D 2 , D 3  and D 5  is relatively small, the loss in D 2 , D 3  and D 5  can be omitted by comparison, and only the selections of D 1  and D 4  are listed in Table 1. It can be seen in  FIG. 6  and Table 1 that the topology according to the embodiments of the present invention has an approximate cost but a higher efficiency as compared with the topology shown in  FIG. 2 . 
       FIG. 7  shows the efficiency comparison between the two forward-flyback converter topologies as shown in  FIG. 2  and  FIG. 3  in the case of load variation (test conditions: input voltage Vin=360V DC, output voltage V 0 =275V DC). It can be seen that, within the variation range from light load to heavy load, the topology according to the embodiment of the present invention maintains high efficiency, and, the efficiency increases with the output power, even up to 96% or more. 
     
       
         
           
               
             
               
                 TABLE 1 
               
               
                   
               
               
                 Selection of Secondary-side Diode Varying with V0 (P0 = 480 W) 
               
               
                   
               
             
            
               
                   
               
            
           
           
               
               
               
               
               
               
               
               
               
               
            
               
                 Topology 
                 Output Current/A 
                 1.5 
                 2 
                 3 
                 4 
                 5 
                 6 
                 8 
                 12 
               
               
                   
                 Output Voltage/V 
                 320 
                 240 
                 160 
                 120 
                 96 
                 80 
                 60 
                 40 
               
               
                 Topology of FIG. 1 
                 Max. Rev. Vol./V 
                 720 
                 544 
                 368 
                 264 
                 224 
                 176 
                 140 
                 88 
               
            
           
           
               
               
               
               
               
               
            
               
                 (D0) 
                 V RRM /V 
                 800 
                 600 
                 400 
                 200 
               
               
                   
                 Forward Voltage 
                 Vf = 1.5 V 
                 Vf = 1.2 V 
                 Vf = 0.9 V 
                 Vf = 0.7 V 
               
            
           
           
               
               
               
               
               
               
               
               
               
               
            
               
                 Topology of FIG. 3 
                 Max. Rev. Vol./V 
                 320 
                 240 
                 160 
                 120 
                 96 
                 80 
                 60 
                 40 
               
            
           
           
               
               
               
               
               
            
               
                 (D1/D2) 
                 V RRM /V 
                 400 
                 200 
                 100 
               
               
                   
                 Forward Voltage 
                 Vf = 0.9 V 
                 Vf = 0.7 V 
                 Vf = 0.5 V 
               
            
           
           
               
               
               
               
               
               
               
               
               
               
            
               
                 Topology of FIG. 2 
                 Max. Rev. Vol./V 
                 320 
                 240 
                 160 
                 120 
                 96 
                 80 
                 60 
                 40 
               
            
           
           
               
               
               
               
               
            
               
                 (D1/D4) 
                 V RRM /V 
                 400 
                 200 
                 100 
               
               
                   
                 Forward Voltage 
                 Vf = 0.9 V 
                 Vf = 0.7 V 
                 Vf = 0.5 V 
               
               
                   
               
            
           
         
       
     
     When the switching power supply is formed by using the converter topology proposed by the present invention, in order to further reduce cost, it may be considered to use a fixed frequency control chip, for example, by using UC3842 as a PWM control chip, a cost-effective solution can be obtained with only a few external components. However, the limitations of the solution are that: the secondary-side resonant circuit cannot operate with high efficiency in a very wide input range, and in order to implement high efficiency, the duty cycle is usually within a range of 0.4 to 0.6, which restricts the input voltage at full load. However, in the case of the application scenario of the Super Charger, since the input voltage range at full load thereof is not wide, the above solution can meet the requirements of the input voltage range while achieving operation at high-efficiency. Therefore, a combination of the topology according to the embodiment of the present invention and the fixed frequency IC is preferably used to constitute the Super Charger of high efficiency and low cost. 
     First Variant 
       FIG. 8  shows the first variant applied to a resonant circuit of the forward-flyback DC-DC converter topology according to the embodiment of the present invention, wherein the resonant circuit adopted is in a form different from that as shown in  FIG. 3 , to implement the zero-current switching of the secondary-side rectifying diode. As shown in  FIG. 8 , the resonant circuit includes a first inductor L S1 , a second inductor L S2 , a first capacitor C 1  and a second capacitor C 2 . The first inductor L S1  and the first capacitor C 1  are connected in series between a first output terminal and an unlike terminal of the forward winding and the flyback winding, the second inductor L S2  and the second capacitor C 2  are connected in series between a second output terminal and the unlike terminal of the forward winding and the flyback winding. 
     Second Variant 
       FIG. 9  shows the second variant applied to a resonant circuit of the forward-flyback DC-DC converter topology according to the embodiment of the present invention, wherein the resonant circuit adopted is in a form different from those as shown in  FIG. 3  and  FIG. 8 , to implement the zero-current switching of the secondary-side rectifying diode. As shown in  FIG. 9 , the resonant circuit includes the first inductor L S1 , the second inductor L S2 , the first capacitor C 1  and the second capacitor C 2 . The first inductor L S1  is connected between the first rectifying switch diode D 1  and the first output terminal, the second inductor L S2  is connected between the second rectifying switch diode D 2  and the second output terminal, the first capacitor C 1  and the second capacitor C 2  are connected in series between the first output terminal and the second output terminal, the unlike terminal of the forward winding and the flyback winding is connected to the intermediate node of the first capacitor C 1  and the second capacitor C 2 . 
     Without departing from the general inventive concept of the present invention, those skilled in the art may think of using other LC resonant circuit to implement the zero-current switching of the secondary-side rectifying diode and of combining it with the exemplary topological structure as shown in  FIG. 4  and other similar topological structure freely, for example, using a resonant circuit having an equivalent circuit same as that of the resonant circuits as shown in  FIG. 3 ,  FIG. 8  and  FIG. 9 . 
     Although the present invention is described by specific embodiments and drawings, the scope of the present invention is not restricted to these specific details. Those skilled in the art will clearly understand that various modifications, substitutions and variations may be made to these details without departing from the spirit and scope of the general inventive concept of the present invention. Therefore, the present invention is not limited to these specific details, exemplary structures and connection manners in a broader sense of embodiment, and the scope thereof is given by the attached claims and their equivalents.