Patent Publication Number: US-9846196-B2

Title: Input current conditioner for precision coulomb counting

Description:
CROSS REFERENCES TO RELATED APPLICATIONS 
     The present application claims the benefit of priority under 35 U.S.C. §119 from U.S. Provisional Patent Application Ser. No. 62/126,833, entitled “Input Current Conditioner for Precision Coulomb Counting,” filed on Mar. 2, 2015, which is hereby incorporated by reference in its entirety for all purposes. 
    
    
     BACKGROUND 
     Technical Field 
     This disclosure generally relates to a power supply system, and more particularly, to tracking coulombs drawn from a power source. 
     Description of Related Art 
     In many applications, it is beneficial to know how much charge has been consumed since a battery was installed or fully charged. This information may be used, for example, for estimating the remaining battery life. Coulomb counters, sometimes referred to as “gas gauges,” are often used in battery-powered circuits to monitor the usage of the battery, which is measured in coulombs (i.e., 1 coulomb=1 ampere*1 second). Graphically, if battery current is plotted on the y-axis vs. time on the x-axis, coulomb usage is represented by the area under the curve. 
     Battery capacity may be rated in units of amp·hour, where 1 amp·hour=3600 amp·sec=3600 coulombs. For example, a battery with a capacity of 1 A·hour, would nominally be expected to deliver 1 A for one hour, 2 A for 0.5 hour, or 0.1 A for 10 hours, etc., before being depleted. 
     Real world loads, however, may not be well-defined. For example, a battery connected to the input of a DC-DC converter may be subject to a load whose input current waveform has AC and DC components. Further, there may be a large dynamic range between the highest and lowest (but nonzero) instantaneous values. 
     In this regard,  FIG. 1  illustrates an example of input current that is provided by a power source (e.g., battery) to a buck DC-DC converter.  FIG. 1  demonstrates how uneven and erratic the input current drawn by a load can be. The current profile may include current spikes  102  due to inductor currents, sequences of which are sometimes referred to as bursts. There may be other components, such as gate charge current spikes  104 . During the active period, there is a DC quiescent current  106 . During the sleep period, there is a second quiescent current  108  that may be lower than the active quiescent current. 
     Accordingly, there is a bursting phase where power is delivered to the load, which is followed by a sleep phase, where power is not delivered to the load but lost nonetheless due to the quiescent sleep current  108 . The burst rate is determined by the load and the output capacitor (C OUT , illustrated in  FIGS. 2A and 2B ). 
     Circuits to count the coulombs at the input of the DC-DC converter presently exist, but they can be large and complicated. Moreover, prior art coulomb counters can be subject to large errors when counting coulombs at low power levels. The greatest source of this error can be the large (but finite) dynamic range of the instantaneous currents. 
       FIG. 2A  illustrates a known approach where a coulomb counter uses a current sense resistor R SENSE    204  in series with the input V IN  to create a small voltage drop across the resistor  204  that is proportional to the input current I BUCK . The profile of the current is illustrated by way of example in waveform  202  (which is a replica of the waveform of  FIG. 1  discussed above. The voltage drop across the sense resistor  204  serves as an input to the coulomb counter  206 , which may include an integrator to calculate and report the area under the current vs. time curve  202 . For example, the LTC2941/2/3 and LTC4150 integrated circuits from Linear Technology Corporation use this technique. 
     Because the voltage drop across the sense resistor  204  represents an efficiency loss, the peak voltage drop across the sense resistor  204  is generally kept small (e.g. 50 mV). Further, the ratio of the highest current to the lowest current may be 100,000 (or even higher), which also prevents using a “large” sense resistor  204  because of the substantial voltage drop across the resistor  204  it would require. The limitation of using a small sense resistor  204  can create an accuracy challenge at small instantaneous currents across the sense resistor because the integrator may have a finite offset that may exceed the IR drop created by the smallest instantaneous input current. Put differently, when the current through the sense resistor  204  is low, this low current cannot be accurately detected due to the low resistance of the sense resistor  204 . 
     Further, at light loads, the DC-DC converter may spend the majority of the time in a sleep state, which results in the highest error of calculating the coulombs drawn from the power source. In addition, due to the AC nature of the input current waveform, bandwidth issues can further affect accuracy. For example, the integrator has a finite bandwidth (i.e., the highest frequency of an input signal at which it can perform the integration with acceptable accuracy). If the input current contains frequency components higher than this bandwidth (e.g., possibly the gate charge current spikes), these additional components typically result in additional error. 
     The burst rate is determined by the load at V OUT  and the output capacitor C OUT . Both the sleep time and the burst time are proportional to C OUT . For example, the sleep and burst times can be calculated by equations 1 and 2 below. 
     
       
         
           
             
               
                 
                   
                     Sleep 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     Time 
                   
                   = 
                   
                     
                       
                         C 
                         OUT 
                       
                       × 
                       
                         V 
                         RIPPLE 
                       
                     
                     
                       I 
                       LOAD 
                     
                   
                 
               
               
                 
                   EQ 
                   . 
                   
                       
                   
                   ⁢ 
                   1 
                 
               
             
             
               
                 
                   
                     Burst 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     Time 
                   
                   = 
                   
                     
                       
                         C 
                         OUT 
                       
                       × 
                       
                         V 
                         RIPPLE 
                       
                     
                     
                       
                         ISW 
                         BUCK 
                       
                       - 
                       
                         I 
                         LOAD 
                       
                     
                   
                 
               
               
                 
                   EQ 
                   . 
                   
                       
                   
                   ⁢ 
                   2 
                 
               
             
           
         
       
     
     Where:
         C OUT  is the output capacitor  220  (e.g., a constant);   ISW BUCK  is the current the buck converter can deliver when in burst mode (e.g., constant for a predetermined set of conditions);   V RIPPLE  is the ripple in the output voltage (e.g., depends on the implementation, but typically small); and   I LOAD  is the load current (e.g., application dependent).       

       FIG. 2B  illustrates an alternate known implementation of counting coulombs. The approach of  FIG. 2B  forgoes the use of the current sense resistor  204  of  FIG. 1 . Instead,  FIG. 2B  includes a coulomb counter  256  that counts only the coulombs associated with inductor current IL when the switch  258  is closed. The implementation of  FIG. 2B  exploits the known shape of the current profile IL  260  flowing in the inductor L based on the particular DC-DC topology in order to simplify the architecture. 
     Thus, instead of determining the area of the uneven and erratic profile of the current  202  of  FIG. 1 , the circuit of  FIG. 2B  essentially takes the area of triangles provided by the current profile  260 . The LTC3335 integrated circuit from Linear Technology Corporation uses this technique. Advantageously, there is no efficiency loss associated with having a separate current sense resistor as in  FIG. 2A . However, some of the current components (such as gate charge current spikes  104  and DC quiescent currents  106  and  108  of  FIG. 1 ) may not be accounted for, resulting in lower (but generally predictable) accuracy at all power levels. 
     A commonality in both prior art implementations is that the raw input current waveform to the coulomb counter may not be “well-behaved” under the various conditions that the circuit may operate and therefore difficult to accurately calculate the coulombs drawn from a power source by a load. 
     Accordingly, prior art approaches may exhibit efficiency loss and may lead to large coulomb counter errors, particularly at light loads. It is with respect to these considerations and others that the present disclosure has been written. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
       The drawings are of illustrative embodiments. They do not illustrate all embodiments. Other embodiments may be used in addition or instead. Details that may be apparent or unnecessary may be omitted to save space or for more effective illustration. Some embodiments may be practiced with additional components or steps and/or without all of the components or steps that are illustrated. When the same numeral appears in different drawings, it refers to the same or like components or steps. 
         FIG. 1  illustrates an example of input current to a buck DC-DC converter. 
         FIGS. 2A and 2B  illustrate prior art coulomb counters used in connection with a DC-DC converter. 
         FIG. 3  illustrates an example of a coulomb counter that uses an input current conditioner and is connected to a buck DC-DC converter. 
         FIG. 4  illustrates an example of a conditioned input current that may flow in the coulomb counter illustrated in  FIG. 3 . 
     
    
    
     DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
     In the following detailed description, numerous specific details are set forth by way of examples in order to provide a thorough understanding of the relevant teachings. However, it should be apparent that the present teachings may be practiced without such details. In other instances, well-known methods, procedures, components, and/or circuitry have been described at a relatively high-level, without detail, in order to avoid unnecessarily obscuring aspects of the present teachings. Some embodiments may be practiced with additional components or steps and/or without all of the components or steps that are described. 
     The methods and circuits disclosed herein generally relate to tracking coulombs drawn from a power source, such as a battery. The coulomb counter counts the coulombs drawn from a power source by a load, such as a DC-DC converter. The coulomb counter includes a reference current source that is configured to regulate a current drawn by the load from the power source. There is a switch coupled between the reference current source and the output of the coulomb counter. The reference current source is configured to provide either a zero or fixed current amount to the load. The opening and closing of the switch is facilitated by a comparator that is configured to compare the voltage at the input of the coulomb counter and its output. 
     By opening and closing the switch, the input current provided to the load is “conditioned” to be a well-defined square wave with only two instantaneous values: a precision trimmed reference current (I REF ) and zero. There is essentially no error associated with the coulomb counter when the input current provided by the reference current is zero, regardless of the duration of the zero. That is because, unlike the prior art implementations that have quiescent current during sleep mode, no coulombs are being drawn from the power source during this period. 
     An example of a “conditioned” input current waveform that may flow in the coulomb counter discussed herein is shown in  FIG. 4 . The duration for which the conditioned input current is ON or OFF is determined by the value of the precision trimmed reference current and the value of a capacitor coupled to the output of the coulomb counter. The average value of the current provided by the conditioned current profile of  FIG. 4  is similar to the average current provided by the current profile of  FIG. 1 . Thus, even though both  FIGS. 1 and 4  provide the same number of coulombs, the conditioned current profile of  FIG. 4  facilitates making a more accurate estimate of the coulombs drawn from the power source. 
     Advantageously, the coulombs drawn from the power source during the time that the input current is at the I REF  level, is simplified to a calculation of I REF *t. Measuring and reporting the coulomb count is as simple as turning a precision trimmed oscillator ON every time the switch is closed, and OFF every time the switch is open, and then counting how many cumulative cycles the oscillator was ON. Significantly, the accuracy of the coulomb counter is no longer based on the DC-DC input current profile of the prior art. Instead, in one aspect, the accuracy may be based on the precision of the control of the DC I REF  and the precision of the time the switch is closed, which are more simple and accurate to implement. 
     With the foregoing overview, reference now is made in detail to the examples illustrated in the accompanying drawings.  FIG. 3  illustrates a coulomb counter  301 , consistent with an exemplary embodiment. By way of example, circuit  300  is coupled to a DC-DC buck converter, although it will be understood that other topologies, such as boost, buck-boost, flyback, etc., are supported as well. Indeed, the circuit and techniques discussed herein are compatible with any load, while they are most advantageous with input current waveforms that have an uneven or erratic profile. 
     The coulomb counter circuit  301  is configured for counting coulombs drawn from the power source  330  (e.g., battery) by a load  315 , configured in the example of  FIG. 3  as a buck DC-DC converter. The coulomb counter  301  includes a reference current source (I REF )  303  coupled between an input node V IN  and an output node V IN2  and configured to regulate a current drawn by the load  315  from the power source  330 . 
     The coulomb counter circuit  301  includes a switch  307  coupled between the reference current source  303  and the output node V IN2  of the coulomb counter  301 . In various embodiments, the switch  307  may be implemented using one or more field effect transistors (e.g., NFET, PFET, an NFET in combination with a PFET, etc.) or their comparable bipolar devices. When the switch is closed, current from the current source I REF    303  is allowed to pass to the output node V IN2 . When the switch  307  is open, current is not allowed to pass to the output node V IN2 . 
     There is a comparator  305  having a first input node (positive terminal) coupled to the input node (V IN ) of the coulomb counter  301 , a second input node (negative terminal) coupled to the output node V IN2  of the coulomb counter  301 , and an output node EN coupled to a control node of the switch  307 . In one embodiment, there is a voltage source  313  coupled between the second input node of the comparator  305  and the output node of the coulomb counter V IN2  and configured to provide a predetermined offset (e.g., 100 mV) for the comparator  305 . The comparator  305  is configured to control an open and closed time for the switch  307 . In one embodiment, the comparator includes hysteresis. The hysteresis effect is discussed in more detail later. 
     In one embodiment, there is an output capacitor C IN2  ( 332 ) coupled to the output node (V IN2 ) of the coulomb counter  301 . The output capacitor  332  is operative to support the load  315  when the switch  307  is open by providing the load  315  a voltage source. The duration for which the conditioned input current is ON or OFF is determined by the value of the current of the reference current source (I REF )  303  and the value of the output capacitor C IN2  ( 332 ). Both the ON time and the OFF time are proportional to C IN2 . The relationship may be better understood in view of equations 3 and 4 below. 
     
       
         
           
             
               
                 
                   
                     
                       IREF 
                       ON 
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     Time 
                   
                   = 
                   
                     
                       
                         C 
                         
                           IN 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           2 
                         
                       
                       × 
                       
                         V 
                         HYST 
                       
                     
                     
                       
                         I 
                         REF 
                       
                       - 
                       
                         I 
                         BUCK 
                       
                     
                   
                 
               
               
                 
                   EQ 
                   . 
                   
                       
                   
                   ⁢ 
                   3 
                 
               
             
             
               
                 
                   
                     
                       IREF 
                       OFF 
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     Time 
                   
                   = 
                   
                     
                       
                         C 
                         
                           IN 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           2 
                         
                       
                       × 
                       
                         V 
                         HYST 
                       
                     
                     
                       I 
                       BUCK 
                     
                   
                 
               
               
                 
                   EQ 
                   . 
                   
                       
                   
                   ⁢ 
                   4 
                 
               
             
           
         
       
     
     Where,
         C IN2  is the output capacitor  332  (e.g., a constant);   I REF  is the reference current  303  (e.g., constant);   V HYST  is the comparator  305  voltage hysteresis (e.g., a constant); and   I BUCK  is the input current to the load during IREF ON  and IREF OFF  (e.g., not a constant).       

     There is an oscillator  309  coupled to the output node of the comparator  305 . The oscillator is enabled when the switch  307  is closed (i.e., the comparator  305  provided an output signal to activate the switch  307 ). The oscillator continues to provide a clock signal at its output (i.e., CLK) as long as the switch is closed. The output of the oscillator is coupled to a counter  311  that receives the clock signal at the output of the oscillator  309  and counts the number of cycles during the time the switch  307  is closed. The count is provided at a second output of the coulomb counter  301 . 
     Thus, the example coulomb counter  301  of  FIG. 3 , conditions the input current I IN  before it provides it to the load (illustrated by way of example as a buck DC-DC converter  315 ). More specifically, the current source  303  may deliver a constant current I REF  (which can be turned ON and OFF with the switch  307 ). In one embodiment, the current source  303  is a low-dropout current source, thereby allowing the output voltage V IN2  to be similar to the input voltage V IN . 
     In the example of  FIG. 3 , the current source  303  is placed between a power source  330  (providing V IN ) and an input to the DC-DC converter  315 , represented by the source of PFET  336  of the load  315 , thereby providing the intermediate output node V IN2 . Additionally, the output voltage (V IN2 ) of the coulomb counter  301  may have a low voltage, low frequency ripple that may be observed at the input to the load  315 , as illustrated in curve  340 . This generally does not cause downstream regulation problems, but it may provide some duty-cycle modulation in the DC-DC converter switch waveform, which can be readily controlled. 
     In one embodiment, if V IN2  is more than a first predetermined threshold (e.g., 150 mV) below V IN  (such as at start up), the switch  307  may turn ON the current source  303  and charge may be delivered from V IN  to V IN2  at the constant rate of I REF . This determination is performed by the comparator  305 . When V IN2  charges up to within a second predetermined threshold voltage (e.g. 100 mV) of V IN , the switch  307  turns OFF the current source I REF    303 , and the load  315  can be supported by a C IN2  capacitor  332  while the current provided by the power source (e.g., battery) goes to zero. 
     In one embodiment, a voltage source  313  may be used to provide a desired voltage offset. For example, the offset may be set at 100 mV, while it will be understood that any suitable offset may be used. Similarly, the first and second predetermined thresholds (e.g., 100 mV and 150 mV) that have been used in  FIG. 3  are example implementations. Accordingly, other suitable first and second predetermined thresholds may be used as well. In one embodiment, the first and second predetermined thresholds are as small as possible, while still allowing I REF  to be accurate. For example, the accuracy of the current source  303  (and by extension the accuracy of the coulomb counter  301 ) generally degrades (or becomes impractical to implement because of the complexity of the additional circuitry required) as the differential voltage across the current source  303  becomes smaller. 
     As noted above, the (e.g., hysteresis) comparator  305  controls the open and closed state of the switch  307  and is configured to sense both the first predetermined threshold and the second predetermined threshold. The output node of the comparator  305  is operative to provide the signal that closes and opens the switch  307  to turn current source I REF    303  ON and OFF, thereby controlling the time that the power source  330  provides current. On average the current provided by the power source  330  is substantially equal to the current provided by the current source  303  (I REF ). The same output node of the comparator  305  that is used to drive the switch  307  may also be used to turn ON and OFF the oscillator  309 . In one embodiment, the oscillator  309  is configured to provide a clock signal (CLK) with a precision period T. This output of the oscillator  309  is used to increment the counter  311 . Accordingly, the counter  311  provides bits at its output node that represent a count n of the coulombs that were drawn from the power source  330  (e.g., battery). 
     Advantageously, the accuracy of the coulomb counter is based now on the precision of the current source  303  providing I REF  and the precision of the period during which the switch is closed (T), which are each substantially controllable and can be configured for a desired precision. For example, in one embodiment, these components only require a trimmed accurate current source and a trimmed accurate time base, both of which are readily available. Accordingly, accuracy is always provided, even at low power levels where prior art techniques typically falter. It is believed that those skilled in the art are adequately familiar with the structure and optimization of current sources and oscillators—such structures are therefore not included for brevity. 
     In one embodiment, the count of the counter  311  may be reset after the power source  330  (e.g., battery) is recharged or replaced. 
     In various embodiments, one or more circuits of the coulomb counter  301  may be powered by the voltage at the output node V IN2 . In one example, all circuits of the coulomb counter  301  are powered by the voltage at the output node V IN2 , thereby ensuring zero power source  330  (e.g., battery) current when the switch  307  has turned off the current source  303 . In this embodiment, any power consumed by the coulomb counter  301  while the switch  307  is open is provided by the output capacitor C IN2    332 . To refresh capacitor C IN2    332  (i.e. bring it back up in voltage), the current source  303  I REF  is turned back ON and any coulombs consumed are counted. Accordingly, the coulomb counter  301  not only counts the coulombs drawn from the battery to power the buck, but also those required to power the coulomb counter  301  itself through bootstrapping. Thus, any charge consumed by the coulomb counter  301  does not manifest itself as a parasitic loss that is not accurately calculated. Rather, the charge consumed by the coulomb counter  301  is attributed as an output “load,” thereby facilitating its accurate calculation by the coulomb counter  301  itself. 
     The components, steps, features, objects, benefits, and advantages that have been discussed herein are merely illustrative. None of them, nor the discussions relating to them, are intended to limit the scope of protection in any way. Numerous other embodiments are also contemplated. These include embodiments that have fewer, additional, and/or different components, steps, features, objects, benefits, and/or advantages. These also include embodiments in which the components and/or steps are arranged and/or ordered differently. 
     For example, any signal discussed herein may be scaled, buffered, scaled and buffered, converted to another mode (e.g., voltage, current, charge, time, etc.,), or converted to another state (e.g., from HIGH to LOW and LOW to HIGH) without materially changing the underlying control method. Although a buck DC-DC converter is used in the examples herein, it will be understood that other topologies, such as boost, buck-boost, flyback, etc., are supported as well. Indeed, the circuit and techniques discussed herein are compatible with any load. 
     In one example, the MOS transistors in  FIG. 3  could be replaced by bipolar transistors. In other embodiments, the circuits could be reconfigured to use PNP transistors instead of NPN transistors (and PMOS transistors instead of NMOS) while still adhering to the principles of the subject matter disclosed herein. Accordingly, it is intended that the invention be limited only in terms of the appended claims. 
     It should be noted that the coulomb counter circuit may place an element in series with the power source  330  (e.g., such as a low-dropout current source  303 ). While this may create an efficiency loss for all loads, the coulombs drawn from the power source  330  are accurately accounted for as discussed above. If the average current source dropout voltage is configured to be 100 mV, this may represent a 2% efficiency loss for a 5V input, or a 3.3% efficiency loss for a 3.3V input. Accordingly, the higher the input voltage from the power source  330 , the lower the percent efficiency loss due to the coulomb counter  301 . 
     Unless otherwise stated, all measurements, values, ratings, positions, magnitudes, sizes, and other specifications that are set forth in this specification, are approximate, not exact. They are intended to have a reasonable range that is consistent with the functions to which they relate and with what is customary in the art to which they pertain. 
     Except as stated immediately above, nothing that has been stated or illustrated is intended or should be interpreted to cause a dedication of any component, step, feature, object, benefit, advantage, or equivalent to the public, regardless of whether it is or is not recited in the claims. 
     All articles, patents, patent applications, and other publications that have been cited in this disclosure are incorporated herein by reference. 
     It will be understood that the terms and expressions used herein have the ordinary meaning as is accorded to such terms and expressions with respect to their corresponding respective areas of inquiry and study except where specific meanings have otherwise been set forth herein. Relational terms such as “first” and “second” and the like may be used solely to distinguish one entity or action from another, without necessarily requiring or implying any actual relationship or order between them. The terms “comprises,” “comprising,” and any other variation thereof when used in connection with a list of elements in the specification or claims are intended to indicate that the list is not exclusive and that other elements may be included. Similarly, an element preceded by an “a” or an “an” does not, without further constraints, preclude the existence of additional elements of the identical type. 
     The Abstract of the Disclosure is provided to allow the reader to quickly ascertain the nature of the technical disclosure. It is submitted with the understanding that it will not be used to interpret or limit the scope or meaning of the claims. In addition, in the foregoing Detailed Description, it can be seen that various features are grouped together in various embodiments for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted as reflecting an intention that the claimed embodiments require more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive subject matter lies in less than all features of a single disclosed embodiment. Thus the following claims are hereby incorporated into the Detailed Description, with each claim standing on its own as a separately claimed subject matter.