Patent Publication Number: US-8971080-B2

Title: Circuit arrangement with a rectifier circuit

Description:
TECHNICAL FIELD 
     Embodiments of the present invention relate to a circuit arrangement with a rectifier. 
     BACKGROUND 
     Rectifiers are electronic circuits or electronic devices that allow a current to flow in a first direction, while preventing a current to flow in an opposite second direction. Such rectifiers are widely used in a variety of electronic circuits in automotive, industrial and consumer applications, in particular in power conversion and drive applications. 
     Conventional rectifiers can be implemented with a diode that conducts a current when forward biased and that blocks when reverse biased. A diode, however, causes relatively high losses when forward biased. These losses are proportional to the current through the diode. In particular in power conversion application or power supply applications in which high current may flow through the rectifier, significant losses may occur. Further, due to reverse recovery effects, a diode (power diode) used in power conversion or drive applications does not immediately block when it changes from the forward biased state to the reverse biased state, so that there may be a time period in which a current flows in the reverse direction. 
     A rectifier can also be implemented with a MOSFET (power MOSFET) and suitable drive circuit for the MOSFET. A conventional power MOSFET includes an integrated diode, known as body diode, that is effective between a drain terminal and a source terminal of the MOSFET. By virtue of this diode a MOSFET always conducts a current when a voltage is applied between the drain and source terminals that reverse biases the MOSFET. In an n-type MOSFET (p-type MOSFET), a voltage reverse biasing the MOSFET is a positive source-drain voltage (negative source-drain voltage). The drive circuit switches the MOSFET on each time the MOSFET is reverse biased. The losses occurring in a MOSFET in the on-state are lower than losses occurring in a diode under similar operating conditions. However, power MOSFETs, that may be used in rectifiers, in drive applications or an power conversion applications, may have a significant output capacitance that needs to be charged/discharged each time the MOSFET is switched on/off. This capacitance causes switching losses and switching delays. 
     There is therefore a general need to provide a circuit arrangement with a rectifier circuit having reduced losses. 
     SUMMARY 
     A first embodiment relates to a circuit arrangement including a rectifier circuit. The rectifier circuit includes a first and a second load terminal, a first semiconductor device having a load path and a control terminal, and a plurality of n, with n&gt;1, second semiconductor devices, each having a load path between a first load terminal and a second load terminal and a control terminal. The second semiconductor devices have their load paths connected in series and connected in series to the load path of the first semiconductor device, with the series circuit with the first semiconductor device and the second semiconductor devices connected between the load terminals of the rectifier circuit. Each of the second semiconductor devices has its control terminal connected to the load terminal of one of the other second semiconductor devices, and one of the second semiconductor devices has its control terminal connected to one of the load terminals of the first semiconductor device. 
     A second embodiment relates to a method of operating a rectifier circuit. The rectifier circuit includes a first and a second load terminal, a first semiconductor device having a load path and a control terminal, and a plurality of n, with n&gt;1, second semiconductor devices, each having a load path between a first load terminal and a second load terminal and a control terminal. The second semiconductor devices have their load paths connected in series and connected in series to the load path of the first semiconductor device, with the series circuit with the first semiconductor device and the second semiconductor devices connected between the load terminals of the rectifier circuit. Each of the second semiconductor devices has its control terminal connected to the load terminal of one of the other second semiconductor devices, and one of the second semiconductor devices has its control terminal connected to one of the load terminals of the first semiconductor device. The method includes detecting an operation parameter of the rectifier circuit, the operation parameter dependent on at least one of a current through the rectifier element in the first semiconductor device, a voltage across the rectifier element, and a voltage between the first load terminal and the second load terminal, and controlling the first semiconductor device to be switched on dependent on the operation parameter. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Examples will now be explained with reference to the drawings. The drawings serve to illustrate the basic principle, so that only aspects necessary for understanding the basic principle are illustrated. The drawings are not to scale. In the drawings the same reference characters denote like features. 
         FIG. 1  schematically illustrates a circuit arrangement with a rectifier circuit; 
         FIG. 2  illustrates a first embodiment of a rectifier circuit including a series circuit with a first semiconductor device and a plurality of second semiconductor devices connected in series; 
         FIG. 3  illustrates a second embodiment of a rectifier circuit including a series circuit with a first semiconductor device and a plurality of second semiconductor devices connected in series; 
         FIG. 4  illustrates a third embodiment of a rectifier circuit including a series circuit with a first semiconductor device and a plurality of second semiconductor devices connected in series; 
         FIG. 5  illustrates an embodiment of a rectifier circuit including a detection circuit and a control drive circuit; 
         FIG. 6  illustrates the rectifier circuit of  FIG. 5  and an embodiment of the control and drive circuit in detail; 
         FIG. 7  that includes  FIGS. 7A and 7B  illustrates embodiments of the detection circuit; 
         FIG. 8  that includes  FIGS. 8A and 8B  illustrates further embodiments of a rectifier circuit including a series circuit with a first semiconductor device and a plurality of second semiconductor devices connected in series; 
         FIG. 9  illustrates a power converter circuit with a boost converter topology; 
         FIG. 10  illustrates a power converter circuit with a buck converter topology; 
         FIG. 11  illustrates a power converter circuit with a flyback converter topology; 
         FIG. 12  illustrates a power converter circuit with a two-transistor-forward (TTF) topology; 
         FIG. 13  illustrates a power converter circuit with a phase-shift zero-voltage-switching (ZVS) full-bridge topology; 
         FIG. 14  illustrates a power converter circuit with a hard switching half-bridge topology; 
         FIG. 15  illustrates a power converter circuit with an LLC resonant DC/DC converter topology; 
         FIG. 16  illustrates a circuit arrangement with a switch and a rectifier circuit according to a further embodiment; 
         FIG. 17  illustrates embodiments of the switch and the rectifier circuit of  FIG. 16 ; 
         FIG. 18  that includes  FIGS. 18A and 18B  illustrates further embodiments of the detection circuit; 
         FIG. 19  illustrates yet another embodiment of the detection circuit; 
         FIG. 20  illustrates an embodiment of a half-bridge including a signal communication path between a low-side control circuit and a high-side rectifier circuit; 
         FIG. 21  that includes  FIGS. 21A to 21C  illustrates a first embodiment of one second semiconductor device implemented as FINFET. 
         FIG. 22  that includes  FIGS. 22A to 22C  illustrates a second embodiment of one second semiconductor device implemented as FINFET. 
         FIG. 23  illustrates a vertical cross sectional view of a semiconductor body according to a first embodiment in which a first semiconductor device and a plurality of second semiconductor devices are implemented in one semiconductor fin. 
         FIG. 24  illustrates a vertical cross sectional view of a semiconductor body according to a second embodiment in which a first semiconductor device and a plurality of second semiconductor devices are implemented in one semiconductor fin. 
         FIG. 25  illustrates a top view of a semiconductor body according to a third embodiment in which a first semiconductor device and a plurality of second semiconductor devices each including several FINFET cells are implemented. 
         FIG. 26  illustrates a vertical cross sectional view of one second semiconductor device including several FINFET cells connected in parallel. 
         FIG. 27  that includes  FIGS. 27A to 27C  illustrates a further embodiment of one second semiconductor device including several FINFET cells connected in parallel. 
         FIG. 28  illustrates two second semiconductor devices of the type illustrated in  FIG. 27  connected in series. 
         FIG. 29  illustrates a vertical cross sectional view of a first transistor according to a further embodiment. 
         FIG. 30  illustrates a vertical cross sectional view of a second transistor according to a further embodiment. 
     
    
    
     DETAILED DESCRIPTION 
     In the following detailed description, reference is made to the accompanying drawings, which form a part thereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. 
       FIG. 1  illustrates a circuit arrangement with a rectifier circuit  10  connected between a first circuit block  201  and a second circuit block  202 . Each of the circuit blocks  201 ,  202  includes at least one of an electronic device, a voltage source, a current source, at least one of a terminal for applying an electrical potential. Some embodiments of the first and second circuit blocks are explained with reference to further figures below. 
     The rectifier circuit  10  includes a first load terminal coupled to the first circuit block  201  and a second load terminal  202  coupled to the second circuit block  202 . The rectifier circuit  10  is configured to conduct a current I 1  when a voltage V 1  between the first and second load terminals  12 ,  13  has a first polarity, and is configured to block when the voltage V 1  has a second polarity opposite the first polarity and has a magnitude that is lower than a voltage blocking capability of the rectifier circuit  10 . The voltage blocking capability defines the maximum voltage that may be blocked by the rectifier circuit  10 . Just for illustration purposes it is assumed that the voltage V 1  has the first polarity when the voltage V 1  is a positive voltage between the first and the second load terminals  12 ,  13 , and that the voltage V 1  has the second polarity when the voltage V 1  is a negative voltage between the first and the second load terminals  12 ,  13 . 
       FIG. 2  illustrates a first embodiment of the rectifier circuit  10 . Referring to  FIG. 2 , the rectifier circuit  10  includes a first semiconductor device  2  and a plurality of second semiconductor devices  3   1 - 3   n . 
     The first semiconductor device  2  has a load path between a first load terminal  22  and a second load terminal  23  and a control terminal  21  and can assume one of an on-state, in which the load path conducts a current, and an off-state, in which the load paths blocks. The first semiconductor device  2  according to  FIG. 1  is implemented as a transistor device (transistor). Specifically, the first semiconductor device according to  FIG. 2  is implemented as a MOSFET where the control terminal  21  is a gate terminal and the first and second  22 ,  23  load terminals are source and drain terminals, respectively. The first semiconductor device will be referred to as first transistor in the following 
     In  FIG. 2  as well as in the following figures reference number “ 3 ” followed by a subscript index denotes the individual second semiconductor devices. Same parts of the individual second semiconductor devices, such as control terminals and load terminals, have the same reference character followed by a subscript index. For example,  3   1  denotes a first one of the second semiconductor devices that has a control terminal  31   1  and first and second load terminals  32   1 ,  33   1 . In the following, when reference is made to an arbitrary one of the second semiconductor devices or to the plurality of the second semiconductor devices, and when no differentiation between individual second semiconductor devices is required, reference numbers  3 ,  31 ,  32 ,  33  without indices will be used to denote the second semiconductor devices and their individual parts. 
     The second semiconductor devices  3  are implemented as transistor devices (transistors) in the embodiment illustrated in  FIG. 5  and will be referred to as second transistors in the following. Each of the second transistors  3  has a control terminal  31  and a load path between a first load terminal  32  and a second load terminal  33 . The load paths  32 - 33  of the second semiconductor devices are connected in series with each other so that the first load terminal of one second transistor is connected to the second load terminal of an adjacent second transistor. Further, the load paths of the second transistors  3  are connected in series with the load path  22 - 23  of the first semiconductor device  2 , so that the first semiconductor device  1  and the plurality of second transistors  3  form a cascode-like circuit. 
     Referring to  FIG. 3 , there are n second transistors  3 , with n&gt;1 (or n≧2). From these n second transistors  3 , a first second transistors  3   1  is the second transistor that is arranged closest to first semiconductor device  2  in the series circuit with the n second transistors  3  and has its load path  32   1 - 33   1  directly connected to the load path  22 - 23  of the first semiconductor device  2 . An n-th second transistors  3 , is the second transistor that is arranged most distant to first semiconductor device  2  in the series circuit with the n second transistors  3 . In the embodiment illustrated in  FIG. 5 , there are n=4 second transistors  3 . However, this is only an example, the number n of second transistors  3  can be selected arbitrarily, namely dependent on a desired voltage blocking capability of the semiconductor device arrangement. This is explained in greater detail herein below. 
     Each of the second transistors  3  has its control terminal  31  connected to one of the load terminals of another one of the second transistors  3  or to one of the load terminals of the first transistor  2 . In the embodiment illustrated in  FIG. 1 , the 1st second transistor  3   1  has its control terminal  31   1  connected to the first load terminal  22  of the first transistor  2 . Each of the other second transistors  3   2 - 3   n−1  have their control terminal  31   2 - 31   n  connected to the first load terminal  32   1 - 32   3  of the second transistor that is adjacent in the series circuit in the direction of the first semiconductor device  2 . Assume, for explanation purposes, that  3   i  is one of the second transistors  3   2 - 3   n  other than the 1st second transistor  3   1 . In this case, the control terminal  31   i  of this second transistor (upper second transistor)  3   i  is connected to the first load terminal  32   i−1  of an adjacent second transistor (lower second transistor)  3   i−1 . The first load terminal  32   i−1  of the lower second transistor  3   i−1  to which the control terminal of the upper second transistor  3   i  is connected to is not directly connected to one of the load terminals  32   i ,  33   i  of this upper second transistor  3   i . According to a further embodiment (not illustrated), a control terminal  31   i  of one second transistor  3   i  is not connected to the first load terminal  31   i−1  of that second transistor  3   i−1  that is directly connected to the second transistor  3   i , but is connected to the load terminal  32   i−k  of a second transistor  3   i−k , with k&gt;1, farther away from the transistor. If, for example, k=2, then the control terminal  31   i  of the second transistor  3   i  is connected to the first load terminal  32   i−2  of the second transistor  3   i−2  that is two second transistors away from the second transistor  3   i  in the direction of the first transistor  2  in the series circuit. 
     Referring to  FIG. 2 , the first transistor  2  and the second transistors  3  can be implemented as MOSFETs. Each of these MOSFETs has a gate terminal as a control terminal  21 ,  31 , a source terminal as a first load terminal  22 , 32 , and a drain terminal as a second load terminal  23 ,  33 . MOSFETs are voltage controlled devices that can be controlled by the voltage applied between the gate and source terminals (the control terminal and the first load terminal). Thus, in the arrangement illustrated in  FIG. 2 , the 1st second transistors  3   1  is controlled through a voltage that corresponds to the load path voltage of the first transistor  2 , and the other second transistors  3   i  are controlled through the load path voltage of at least one second transistor  3   i−1  or  3   i−2 . The “load path” voltage of one MOSFET is the voltage between the first and second load terminal (drain and source terminal) of this MOSFET. 
     In the embodiment illustrated in  FIG. 2 , the first transistor  2  is a normally-off (enhancement) transistor, while the second transistors  3  are normally-on (depletion) transistors. However, this is only an example. Each of the first semiconductor device  2  and the second transistors  3  can be implemented as a normally-on transistor or as a normally-off transistor. The individual transistors can be implemented as n-type transistors or as p-type transistors. It is even possible to implement the first transistor  2  as a transistor of a first conduction type (n-type or p-type) and to implement the second transistors as transistors of a second conduction type (p-type or n-type) complementary to the first type. 
     Implementing the first transistor  2  and the second transistors  3  as MOSFETs is only an example. Any type of transistor can be used to implement the first semiconductor device  2  and the second transistors  3 , such as a MOSFET, a MISFET, a MESFET, an IGBT, a JFET, a FINFET, a nanotube device, an HEMT, etc. Independent of the type of device used to implement the first semiconductor device  2  and the second semiconductor devices  3 , these devices are connected such that each of the second semiconductor devices  3  is controlled by the load path voltage of at least one other second semiconductor devices  3  or the first semiconductor device  2  in the series circuit. 
     The semiconductor device arrangement  1  with the first transistor  2 , and the second transistors  3  can be switched on and off like a conventional transistor by applying a suitable drive voltage or drive signal S 2  to the first semiconductor device  2 . The control terminal  21  of the first transistor  2  forms a control terminal  11  of the overall arrangement  1 , and the first load terminal  21  of the first transistor  2  and the second load terminal of the n-th second transistor  3   n  form the first and second load terminals  12 ,  13 , respectively, of the overall arrangement. The drive signal S 2  for switching on and off the first transistor  2  and, therefore, the semiconductor device arrangement, can be generated in different ways explained below. When the first transistor  2  is switched on, the semiconductor device arrangement  1  may conduct a current in both directions, namely the first direction and the second direction explained with reference to  FIG. 1 . However, the drive signal S 2  is generated such that it switches on the semiconductor device arrangement  1  only when the voltage V 1  between the first and second load terminals  12 ,  13  has the first polarity. That is, when the voltage V 1  is a positive voltage between the first and second load terminals in the embodiment of  FIG. 2 . Thus, the semiconductor device arrangement  1 , acts as a rectifier element in the rectifier circuit  10 . 
     The operating principle of the semiconductor device arrangement  1  is explained in the following. Just for explanation purposes it is assumed that the first transistor  2  is implemented as an n-type enhancement MOSFET, that the second transistors  3  are implemented as n-type depletion MOSFETs or n-type JFETs, and that the individual devices  2 ,  3  are interconnected as illustrated in  FIG. 5 . The basic operating principle, however, also applies to semiconductor device arrangements implemented with other types of first and second semiconductor devices. 
     It is commonly known that depletion MOSFETs or JFETs, that can be used to implement the second transistors  3 , are semiconductor devices that are in an on-state when a drive voltage (gate-source voltage) of about zero is applied, while depletion MOSFETs or JFETs are in an off-state when the absolute value of the drive voltage is higher than a pinch-off voltage of the device. The “drive voltage” is the voltage between the gate terminal and the source terminal of the device. In an n-type depletion MOSFET or JFET the pinch-off voltage is a negative voltage, while the pinch-off voltage is a positive voltage in a p-type depletion MOSFET or JFET. 
     When a voltage is applied between the first and second load terminals  12 ,  13  and when the first transistor  2  is switched on by applying a suitable drive potential (drive signal) S 2  to the control terminal  11 , the 1st second transistor  3   1  is conducting (in an on-state), the absolute value of the voltage across the load path  22 - 23  of the first transistor  2  is too low so as to pinch-off the 1st second transistor  3   1 . Consequently, the second transistor  3   2  controlled by the load path voltage of second transistor  3   1  is also starting to conduct, etc. In other words, the first transistor  2  and each of the second transistors  3  are finally conducting so that the semiconductor arrangement  1  is in an on-state. 
     The first transistor  1  implemented as a MOSFET may be implemented with an internal diode D 2  (that is also illustrated in  FIG. 2 ) known as body diode. The body diode is parallel to the load path of the transistor. In an n-type MOSFET (as illustrated in  FIG. 2 ) an anode terminal of the diode D 2  corresponds to the source terminal  22  of the MOSFET, while a cathode terminal corresponds to the drain terminal  23 . Thus, a positive source-drain voltage (negative drain-source voltage) of the first transistor  1  forward biases the body diode D 2 . In a p-type MOSFET a negative source-drain voltage (positive drain-source voltage) forward biases the body diode. 
     Referring to  FIG. 2 , the first transistor  1  is connected such that a load path voltage V 1  with the first polarity (as illustrated in  FIG. 2 ) forward biases the body diode D 2 . When the body diode D 2  is forward biased, a voltage drop across the body diode D 2  switches on the 1st second transistor  3   1 , which again switches on the 2nd second transistor  3   2 , and so on. Thus, when the first transistor  1  is switched off, the semiconductor device arrangement by virtue of the body diode D 2  automatically operates as a rectifier element that conducts a current when the load path voltage V 2  has the first polarity. When the polarity of the external voltage V 1  changes to the second polarity (which is opposite to the polarity illustrated in  FIG. 2 ), the body diode D 2  is reverse biased so that the 1st second transistor  3   1  starts to switch off when the absolute value of the load-path voltage reaches the pinch-off voltage of the 1st second transistor  3   1 . 
     When the 1st second transistor  3   1  is switched off, the voltage drop across its load path increases so that the 2nd second transistor  3   2  is switched off, which in turn switches off the 3rd second transistor, and so on, until each of the second transistors  3  is switched off and the semiconductor device arrangement  1  is finally in a stable off-state. The external voltage V 1  with the second polarity applied between the second and first terminals  13  and  12  switches as many 2nd transistors from the on-state to the off-state as required to distribute the external voltage over the first semiconductor device  2  and the second transistors  3 . When applying a low external voltage V 1  with the second polarity, some second transistor  3  are still in the on-state, while others are in the off-state. The number of second transistors  3  that are in the off-state increases as the external voltage V 1  with the second polarity increases. Thus, when a high external voltage V 1  with the second polarity is applied, that is in the range of the voltage blocking capability of the overall semiconductor device arrangement  1 , the first semiconductor device  1  and each of the second transistors  3  are in the off-state 
     When the semiconductor device arrangement  1  is in an off-state and when the external voltage V 1  changes the polarity to the first polarity. As soon as the voltage across the body diode D 2  drops to a voltage of about zero, the normally-on 1st second transistors  3   1  switches on which in turn switches on the 2nd second transistor  3   2 , and so on. This continues until each of the second transistors  3  is again switched on. The body diode D 2  conducts as soon as the voltage V 1  with the first polarity increases to the forward voltage of the body diode D 2 . This forward voltage is about 0.7V when the body diode (and the other semiconductor devices) is implemented in silicon. 
     Although the body diode D 2  enables a current flow in the first direction when the load voltage V 1  has the first polarity, the first transistor  1  through the drive signal  2  may additionally be switched on when the voltage V 1  has the first polarity in order to reduce losses. Losses occurring in the body diode D 2  correspond to the product of forward voltage of the diode, which is about 0.7V when the first transistor  1  is implemented in silicon technology, and the current I 1 . This voltage drop across the body diode D 2  may be reduced to below the forward voltage when switching on the first transistor  1 . When the first transistor  1  is in the on-state (switched on) the body diode D 2  is bypassed. When the first transistor  1  is switched off and the external voltage V 1  still has the first polarity, the body diode D 2  takes the current and keeps the second transistor  3  switched on until the external voltage changes to the second polarity. 
     It is desirable to switch off the first transistor  1  before the voltage V 1  changes to the second polarity in order to prevent a current flow in the second direction. Embodiments of drive circuits and drive schemes that switch on the first transistor  1  only when the voltage V 1  has the first polarity are explained below. 
     Switching states of the second transistors  3  connected in series with the first transistor  2  are dependent on the switching state of the first transistor  2  and follow the switching state of the first transistor  2  when the voltage V 1  has the second polarity. Thus, the second transistors  3  are switched off when the first transistor  2  is switched off and when the voltage V 1  has the second polarity. Further, by virtue of the body diode D 2  the second transistors  3  are switched on independent of the switching state of the first transistor  1  when the voltage V 1  has the first polarity. In this case, switching on the first transistor  1  helps to reduce the losses. 
     In the following, an “on-state” of the semiconductor device arrangement (rectifier element)  1  is an operation state in which the voltage V 1  has the first polarity and in which the first transistor  1  is switched on. An “off-state” is an operation state in which the voltage V 1  has the second polarity and the first transistor  1  is switched off. The semiconductor arrangement  1  has a low resistance between the first and second load terminals  12 ,  13  in the on-state, and has a high resistance between the first and second load terminals  12 ,  13  in the off-state. In the on-state, an ohmic resistance between the first and second load terminals  12 ,  13  corresponds to the sum of the on-resistances R ON  of the first semiconductor device  2  and the second transistors  3  (where the on-resistance is slightly increased when the first transistor  1  is switched off and the body diode D 2  conducts the current). A voltage blocking capability, which is the maximum voltage that can be applied between the first and second load terminals  12 ,  13  when the semiconductor arrangement is in an off-state before an Avalanche breakthrough sets in, corresponds to the sum of the voltage blocking capabilities of the first transistor  2  and the second transistors  3 . The first transistor  1  and the individual second transistors may have relatively low voltage blocking capabilities, such as voltage blocking capabilities of between 3V and 50V. However, dependent on the number n of second transistors  3  a high overall voltage blocking capability of up to several 100V, such as 600V or more, can be obtained. 
     The voltage blocking capability and the on-resistance of the semiconductor arrangement  1  are defined by the voltage blocking capabilities of the first transistor  2  and the second transistors  3  and by the on-resistances of the first transistor  2  and the second transistors  3 , respectively. When significantly more than two second transistors are implemented (n&gt;&gt;2), such as more than 5, more than 10, or even more than 20 second transistors  3  are implemented, the voltage blocking capability and the on-resistance of the semiconductor arrangement  1  are mainly defined by the arrangement  30  with the second transistors  3 . The overall semiconductor arrangement  1  can be operated like a conventional power transistor, where in a conventional power transistor, an integrated drift region mainly defines the on-resistance and the voltage blocking capability. Thus, the arrangement  30  with the second transistors  3  has a function that is equivalent to the drift region in a conventional power transistor. The arrangement  30  with the second transistors  30  will, therefore, be referred to as active drift region (ADR) or active drift zone (ADZ). The overall semiconductor device arrangement  1  of  FIG. 2  can be referred to as ADZ transistor or ADR transistor (ADZ transistor) or as ADRFET (ADZFET), when the first semiconductor device is implemented as a MOSFET. 
     When the semiconductor device arrangement  1  is in the off-state, the voltage V 1  (with the second polarity) applied between the first and second load terminals  12 ,  13  is distributed such that a part of this voltage drops across the load path  22 - 23  of the first transistor  2 , while other parts of this voltage drop across the load paths of the second transistors  3 . However, there may be cases in which there is no equal distribution of this voltage to the second transistors  3 . Instead, those second transistors  3  that are closer to the first semiconductor device  2  may have a higher voltage load than those second transistors  3  that are more distant to the first semiconductor device  2 . 
     In order to more equally distribute the voltage to the second transistors  3 , the semiconductor arrangement optionally includes voltage limiting means  7   1 - 7   n  that are configured to limit or clamp the voltage across the load paths of the second transistors  3 . Optionally, a clamping element  7   0  is also connected in parallel to the load path (between the source and drain terminals) of the first semiconductor device  2 . These voltage clamping means  7   0 - 7   n  can be implemented in many different ways. Just for illustration purposes the clamping means  7   0 - 7   n  illustrated in  FIG. 2  include Zener diodes  7   0 - 7   n , with each Zener diode  7   0 - 7   n  being connected in parallel with the load path of one of the second transistors  3  and, optionally, the first transistor  2 . 
     Instead of the Zener diodes  7   0 - 7   n , tunnel diodes, PIN diodes, avalanche diodes, or the like, may be used as well. According to a further embodiment (not illustrated), the individual clamping elements  7   0 - 7   n  are implemented as transistors, such as, for example, p-type MOSFETs when the second transistors  3  are n-type MOSFETs. Each of these clamping MOSFETs has its gate terminal connected to its drain terminal, and the load path (the drain-source path) of each MOSFET is connected in parallel with the load path of one second transistor  3 . 
     The individual clamping elements, such as the Zener diodes  7   0 - 7   n  illustrated in  FIG. 2  can be integrated in the same semiconductor body as the first transistor  2  and the second transistors  3 . However, these clamping elements could also be implemented as external devices arranged outside the semiconductor body. 
     As compared to a conventional power transistor with an integrated body diode, the semiconductor device arrangement  1  with the first transistor  2  and the plurality of second transistors  3  has reduced switching losses and switches faster from the off-state to the on-state. In a conventional power transistor, switching losses occur by charging an output capacitance of the transistor at the time of switching on and by discharging the output capacitance at the time of switching off. The output capacitance (C OSS ) includes an internal drain-source capacitance (C DS ) and an internal gate-drain capacitance (C GD ) of the transistor. Losses further occur due to reverse recovery effects in the body diode. When the body diode is forward biased, electrical charges are stored in the body diode. These charges have to be removed when the body diode is reverse biased before the body diode blocks. Storing charges in the body diode and removing charges from the body diode induces losses. These losses increase with the amount of charges stored in the forward biased body diode, where this amount increases as the voltage blocking capability of the power transistor increases. 
     In the semiconductor device arrangement (ADRFET)  1  the output capacitance of the first transistor  2 , that may have a voltage blocking capability of several volts up to several 10V, is lower than the output capacitance of a conventional power transistor, that may have a voltage blocking capability of up to several 100V. Further, less charges are stored in the body diode of the first transistor  2  when the body diode D 2  is forward biased. Thus, losses occurring in the first transistor  2  of the ADRFET  1  are lower than losses occurring in a power MOSFET having the same voltage capability of the ADRFET  1 . The low output capacitance of the first transistor  2  not only keeps switching losses low, but also results in high switching speeds, which means in fast transitions between the on-state and the off-state of the switch  1 , and vice versa. 
     Gate-source capacitances, gate-drain capacitances and drain source capacitances of the second transistors  3  are also charged and discharged when the switch  1  is switched on and off. However, electrical charges required for charging these capacitances of the second transistors  3  are mainly kept in the arrangement  30  with the second transistors  3 , so that these charges do not have to be provided by the drive circuit  20  in each switching process. These charges are provided via the load path of the ADRFET. Further, by virtue of the relatively low voltage blocking capabilities of the second transistors  3 , the sum of these capacitances of the second transistors  3  is lower than the corresponding output capacitance of a power transistor having the same voltage blocking capability as the ADRFET  1 . 
       FIG. 3  illustrates a further embodiment for implementing the rectifier element (ADRFET)  1  of the rectifier circuit  10 . In the rectifier element  1  of  FIG. 3  the first transistor  2  is implemented with a depletion MOSFET, specifically with an n-type depletion MOSFET. Like in the embodiment of  FIG. 2 , the second transistors  3  of  FIG. 3  may be implemented as depletion transistors, specifically as n-type depletion transistors. The arrangement  30  with the second transistor is only schematically illustrated in  FIG. 3 . The individual second transistors of the arrangement  30  may be interconnected as explained with reference to  FIG. 2 . The operating principle of the rectifier element  1  of  FIG. 3  corresponds to the operating principle of the rectifier element of  FIG. 2  with the difference that a negative drive voltage (gate-source voltage) is required to switch off the first transistor  2  of  FIG. 3 , while the enhancement transistor  2  of  FIG. 2  already switches when the gate-source voltage decreases below a positive threshold voltage. 
     Referring to the explanation above, the first transistor  2  of the rectifier element  1  receives a drive signal S 2 . According to one embodiment, the drive signal S 2  is generated such that it switches the first transistor  2  on when the external voltage V 1  has the first polarity and switched the first transistor  2  off when the external voltage has the second polarity. According to one embodiment, the drive signal S 2  is an externally generated drive signal or is dependent on such externally generated drive signal. An externally generated drive signal is a drive signal generated by an external circuit and is provided to the rectifier circuit  10 . According to a further embodiment, the drive signal S 2  is an internally generated drive signal. An internally generated drive signal is a drive signal generated in the rectifier circuit  10 . 
       FIG. 4  schematically illustrates an embodiment of the rectifier circuit  10  that receives an externally generated drive signal Sin. According to one embodiment, the externally generated drive signal Sin is provided to the first transistor  2  as the drive signal S 2  of the transistor  2 . According to a further embodiment, a drive circuit  14  (illustrated in dashed lines) receives the externally generated drive signal Sin and generates the drive signal S 2  of the transistor  2  from the received drive signal Sin. The drive circuit  14  may be configured to adapt signal levels of the received drive signal Sin such that signal levels suitable for driving the first transistor  2  are obtained. 
     The rectifier element  1  of  FIG. 4  corresponds to the rectifier element of  FIG. 2 . However, this is only an example. The rectifier element  1  could be implemented like any of the rectifier elements explained before. 
       FIG. 5  illustrates an embodiment of a rectifier circuit  10  in which a drive signal S 2  of the first transistor  2  is internally generated. Referring to  FIG. 5 , the rectifier circuit  10  includes a control and drive circuit  8  and a detection circuit  9 . The control and drive circuit  8  receives a detection signal S D  from the detection circuit  9  and is configured to generate the drive signal S 2  dependent on the detection signal S D . The detection circuit  9  is configured detect (evaluate) an operation parameter of the rectifier circuit. The operation parameter is dependent on at least one of a current through the rectifier element (body diode) D 2  in the first semiconductor device  2 , a voltage across the rectifier element D 2 , and a voltage between the first load terminal  12  and the second load terminal  13 . 
     According to one embodiment, the detection circuit  9  provides as the detection signal S D  a current measurement signal representing the current I 1 . In this case, the detection signal S D  includes an information on the current direction (corresponding to the sign of the detection signal S D ) and an information on the magnitude of the current I 1 . In this embodiment, the control and drive circuit  8  may be configured to switch on the first transistor  2  each time the detection signal S D  indicates that the current I 1  flows in the first direction (which in the embodiment of  FIG. 5  is the current flow direction illustrated in  FIG. 5 ). The body diode D 2  of the first transistor  2  enables a current flow in the first direction I 1  before the first transistor  2  is switched on. The first transistor  2  may be switched off when the current I 1  falls below a predefined current threshold. A decrease of the current I 1  to below the current threshold may indicate that the current I 1  is probably about to decrease to zero and that a polarity of the voltage V 1  is probably about to change to the second polarity (the polarity opposite to the polarity illustrated in  FIG. 5 ). 
     According to a further embodiment, the detection circuit  9  provides as the detection signal S D  a current measurement signal representing the current I 1  and the control and drive circuit  8  is configured to determine a time variation of the current measurement signal S D . According to one embodiment, the control and drive circuit  8  is configured to switch on the first transistor  2 , when the detection circuit S D  indicates that the current I 1  flows in the first direction. Further, the control and drive circuit  8  is configured to switch off the first transistor  2  when the current I 1  flowing in the first direction decreases and when a slope of the (decreasing) current is higher than a predefined falling slope threshold. This is equivalent to the fact that a (negative) differential coefficient (dI 1 /dt) of the current I 1  has a magnitude higher than the predefined slope threshold. Alternatively, the control and drive circuit  8  switches on the first transistor  2  when the current I 1  flows in the first direction and increases and when the slope of the increasing current I 1  is above a further slope threshold. This is equivalent to the fact that the positive differential coefficient (dI 1 /dt) of the current I 1  is above the further slope threshold. 
     According to yet another embodiment, the detection signal S D  represents a voltage V 2  across the body diode. The polarity of this voltage V 2  corresponds to the polarity of the voltage V 1  between the load terminals  12 ,  13 . The body diode voltage V 2  has the first polarity when it forward biases the body diode D 2  and has the second polarity when it reverse biases the body diode. The body diode D 2  starts to conduct, when the voltage V 2  has the first polarity and a magnitude corresponding to the forward voltage of the body diode D 2  (about 0.7V in silicon). According to one embodiment, the control and drive circuit  8  is operable to switch on the first transistor  1  when the detection signal S D  indicates that the body diode voltage V 2  has the first polarity and reaches a first voltage threshold. The first voltage threshold may be below the forward voltage of the body diode D 2 . In this case the control drive circuit  8  may switch on the first transistor  2  before the body diode conducts. However, due to the propagation delays the body diode voltage may increase to the forward voltage between the time when the body diode voltage V 2  reaches the first voltage threshold and the time when the first transistor  1  switches on, so that the body diode D 2  is conducting before the first transistor  1  switches on. The control and drive circuit  8  may further be operable to switch off the first transistor  1 , when the detection signal indicates that the body diode voltage V 2  has the first polarity and falls to a second voltage threshold, such as zero. 
     According to a further embodiment in which the detection signal S D  represents the body diode voltage V 2 , the control and drive circuit  8  is operable to switch on the first transistor  1  when the detection signal S D  indicates that the body diode voltage V 2  has the first polarity and increases and that a slope of the increasing voltage reaches a predefined first voltage slope threshold. Further, the control and drive circuit  8  is operable to switch off the first transistor  1  when the detection signal S D  indicates that the body diode voltage V 2  has the first polarity and decreases and that a slope of the decreasing voltage reaches a predefined second voltage slope threshold. The control and drive circuit  8  may differentiate (calculate a time derivative) of the detection signal S D  in order to obtain the slopes of rising and falling edges of the body diode voltage V 2 . 
       FIG. 6  illustrates one embodiment of the control and drive circuit  8  in greater detail. In the embodiment of  FIG. 6 , the detection circuit  9  is implemented as a current sensor that is configured to measure the current I 1  through the rectifier element  1  and that generates a current measurement signal S D  as the detection signal. The control and drive circuit  8  includes a supply circuit  81  configured to provide a supply voltage V SUP  and an evaluation and drive circuit  82 . The evaluation and drive circuit  82  receives the supply voltage V SUP  and the detection circuit S D  and is configured to generate the drive signal S 2  from the supply voltage V SUP  dependent on the detection signal S D . The evaluation and drive circuit  82  may be configured to evaluate the detection signal S D  as explained in connection with  FIG. 5  and to generate the drive signal S 2  dependent on the evaluation. 
     The supply circuit  81  of  FIG. 6  includes a capacitive storage element  183 , and a rectifier element  181 , such as a diode, connected in series with the capacitive storage element  183 . The series circuit with the capacitive storage element  183  and the rectifier element  181  is connected between the load terminals  13 ,  12  of the rectifier element  1 . The capacitive storage element  183  is charged each time the voltage V 1  across the rectifier element  1  has the second polarity, which is when the first transistor  2  is to be switched off. The rectifier element  181  prevents the capacitive storage element  183  from being discharged when the voltage V 1  changes to the first polarity. Optionally, the supply circuit  81  further includes a voltage limiting element that is configured to limit the voltage across the capacitive storage element  183 . According to one embodiment, the voltage limiting element  182  is implemented as a depletion MOSFET or a JFET and is connected in series with the capacitive storage element  183 . The capacitive storage element  183  is connected between the source terminal and the gate terminal of the depletion MOSFET (JFET). The depletion MOSFET (JFET) pinches off when the voltage across the capacitive storage element  183  equals the pinch-off voltage of the depletion MOSFET (JFET). This pinch-off voltage is selected such that the supply voltage V SUP  reaches a predefined voltage, such as, e.g., 15V, 10V, 5V or the like. Implementing the voltage limiting element  182  as a depletion MOSFET or JFET is only an example. Any other type of voltage limiting element may be used as well. 
       FIG. 7A  illustrates one embodiment of the current sensor  9  of  FIG. 6 . Referring to  FIG. 7 , the current sensor includes a current mirror with a first current mirror transistor  91   1  and a second current mirror transistor  91   2 . The two current mirror transistors  91   1 ,  91   2  have their control terminals (gate terminals) connected, and a load path (drain-source path) of the first current mirror transistor  91   1  is connected in series with the load path of the first transistor  2 . The first current mirror transistor  91   1  is connected in series with a first resistor  91   6 , with the series circuit with the first current mirror transistor  91   1  and the first resistor  91   6  being connected between the first transistor  2  and the transistor arrangement  30 . The first current mirror transistor  91   1  is connected as a diode and has its control terminal (gate terminal) connected with one of its load terminals (drain terminal). The second current mirror transistor  91   2  has its load path connected in series with a second resistor  91   5  and a further transistor  91   3 , with this series circuit being connected between the first load terminal  12  and the transistor arrangement  30 . 
     In the embodiment of  FIG. 7 , the current mirror transistors  91   1 ,  91   2  are implemented as MOSFETs, in particular as p-type MOSFET, which each have their source terminal coupled to the arrangement  30  with the second transistors, via the first resistor  91   6  and the second resistor  91   5 , respectively. The further transistor  91   3  is of the same type as the first transistor  2  and has its load path connected between the first load terminal  12  and the second current mirror transistor  91   2 . The further transistor  91   3  receives the drive signal S 2  and is switched on and off synchronously with the first transistor  2 . The further transistor  91   3  also includes a body diode. However, this body diode is not explicitly illustrated in  FIG. 7A . 
     Referring to  FIG. 7A , the detection circuit  9  further includes an amplifier, such as an operation amplifier (OA). The amplifier receives a voltage across the second resistor  91   5  as an input signal and provides the detection signal S D . The detection signal S D  represents the amplitude of the current I 1  through the first transistor  2  (including the body diode D 2 ). 
       FIG. 7B  illustrates a further embodiment of a detection circuit  9 . The detection circuit of  FIG. 7B  is a modification of the detection circuit of  FIG. 7A  and further includes a second current mirror with a third current mirror transistor  91   7  connected as a diode and a fourth current mirror transistor  91   8 . These two current mirror transistors  91   7 ,  91   8  have their control terminals (gate terminals) connected together. The second current mirror is connected between the first current mirror and the first transistor  2  and the further transistor  91   3 , where the load path of the third current mirror transistor  91   7  is connected between the first current mirror transistor  91   1  and the first transistor  2 , and the load path of the fourth current mirror transistor  91   8  is connected between the second current mirror transistor  91   2  and the further transistor  91   3 . The detection signal S D  is again available at the output of the amplifier  91   4 . While the detection circuit  9  of  FIG. 7A  is only capable of measuring the current I 1  when the current I 1  has the first direction (as indicated in  FIG. 7A ), the detection circuit  9  of  FIG. 7B  is capable of measuring the current I 1  in both directions. 
       FIG. 8A  illustrates a further embodiment of the rectifier circuit  10 . In this embodiment, the first semiconductor element  2  of the rectifier element  1  is implemented as a diode. The operating principle of this diode  2  corresponds to the operating principle of the body diode D 2  of the first transistor in the rectifier elements  1  explained before. The diode  2  of  FIG. 8  may be implemented as the body diode of a MOSFET that has its gate terminal connected to its source terminal. That is, a gate terminal of the MOSFET is not connected to a drive circuit or the like. 
     The operating principle of the rectifier circuit  10  of  FIG. 8A  corresponds to the operating principle of the rectifier circuit  10  of  FIG. 2 , when the body diode D 2  of the first transistor  2  of  FIG. 2  is conducting. The rectifier circuit  10  of  FIG. 8  with the first semiconductor element  2  implemented as a diode has higher losses than a rectifier circuit  10  with the first semiconductor device  2  implemented as a transistor. However, the losses of the rectifier element  1  with the diode  2  and the arrangement  30  with the plurality of second transistors has lower losses and switches off faster than a conventional diode having the same voltage blocking capability as the rectifier circuit  10 . 
       FIG. 8B  illustrates another embodiment of a the rectifier circuit. In this embodiment, the first semiconductor device  2  is implemented with a p-type transistor, specifically a p-type MOSFET, This transistor is connected as a diode and has its control terminal (gate terminal) connected with one of its load terminals (drain terminal). In the embodiment of  FIG. 8B , the source terminal of the MOSFET is connected to the first load terminal  12 , while the drain terminal is connected to the transistor arrangement  30 . The transistor arrangement may be implemented as explained with reference to  FIG. 2  before. In particular, the transistor arrangement  30  may be implemented with n-type depletion MOSFETs or JFETs. 
     The rectifier arrangement of  FIG. 8B  conducts a current I 1  in the first direction (the direction indicated in  FIG. 8B ) when the voltage V 1  between the load terminals  12 ,  13  has the first polarity, so that a voltage V 2  across the MOSFET  2 , has the first polarity, and when the voltage V 2  across the transistor reaches the threshold voltage of the MOSFET  2 . According to one embodiment, the MOSFET is implemented with a threshold voltage of about 0V. 
     The rectifier circuit  10  as explained before may be implemented in a variety of circuit applications, such as industrial, automotive or consumer electronic applications. In particular, the rectifier circuit  10  may be used in power converter circuits operable to generate an output voltage from an input voltage. Embodiments of some power converter circuits including at least one rectifier circuit  10  of the type explained before are explained with reference to drawings below. 
       FIG. 9  illustrates an embodiment of a power converter circuit with a boost converter topology. Referring to  FIG. 9 , the converter circuit includes input terminals  201 ,  202  for receiving an input voltage Vin and output terminals  203 ,  204  for providing an output voltage Vout. An inductive storage element  205 , such as a choke, is connected in series with a switch  206 . The series circuit with the inductive storage element  205  and the switch  206  is connected between the input terminals  201 ,  202 . A series circuit with a rectifier circuit  10  and a capacitive storage element  207  is connected in parallel with the switch  206 , where the output voltage Vout is available across the capacitive storage element  207 . The rectifier circuit  10  may be implemented in accordance with one of the embodiments explained before. 
     Referring to  FIG. 9 , the power converter circuit further includes a drive circuit  208  that is configured to provide a pulse-width modulated (PWM) drive signal S 206  to the switch  206  dependent on an output signal Sout. The output signal Sout is dependent on the output voltage Vout and represents the output voltage Vout. The drive circuit  208  may be implemented like a conventional PWM controller and is configured to adjust a duty-cycle of the drive signal S 206  such that the output voltage Vout equals a pre-defined set voltage. 
     The operating principle of the power converter circuit of  FIG. 9  is as follows: Each time the switch  206  is switched on, energy is magnetically stored in the inductive storage element  205 . When the switch  206  is switched off, a current I 1  through the inductive storage element  205  continuous to flow, where this current flows through the rectifier circuit  10  to the output terminals  203 ,  204  and the capacitive storage element  207 , respectively. The output voltage Vout is a DC voltage. The input voltage Vin may be a DC voltage or an AC voltage. The output voltage Vout is higher than the input voltage Vin or higher than an amplitude of an input voltage Vin. 
     According to one embodiment, the rectifier circuit  10  is operable to receive an external drive signal Sin. This external drive signal Sin may be provided by the control circuit  208 . In this embodiment, the control circuit  208  may be implemented such that it switches on the first transistor in the rectifier circuit  10  each time the switch  206  is switched off, and switches off the first transistor each time the switch  206  is switched or each time the current I 1  decreases to zero. However it is also possible to implement the rectifier circuit  10  (and each of the rectifier circuits explained below) such that a drive signal for the first transistor  2  (not illustrated in  FIG. 9 ) is internally generated, as explained with reference to  FIGS. 5 and 6 , or such that the rectifier circuit  10  is implemented with a diode as the first semiconductor element, as explained with reference to  FIG. 8 . 
       FIG. 10  illustrates an embodiment of a power converter circuit with a buck converter topology. In this embodiment, a series circuit with a switch  306 , an inductive storage element  305  and a capacitive storage element  307  is connected between input terminals  301 ,  302 . The input terminals  301 ,  302  are operable to receive an input voltage Vin. An output voltage Vout is available between output terminals  303 ,  304  across the capacitive storage element  307 . A rectifier circuit  10  is connected in parallel with the series circuit with the inductive storage element  305  and the capacitive storage element  307 . The rectifier circuit  310  may be implemented in accordance with one of the embodiments explained before. 
     Referring to  FIG. 10 , a control circuit  308  generates a drive signal S 306  for the switch  306 . The drive signal is a pulse-width modulated (PWM) drive signal generated by the control circuit  308  dependent on an output signal Sout. The output signal Sout represents the output voltage Vout. The control circuit  308  adjusts the duty-cycle of the drive signal S 306  such that the output voltage Vout corresponds to a pre-defined set voltage. 
     The operating principle of the power converter circuit of  FIG. 10  is as follows: Each time the switch  306  is switched on, a current I 1  flows driven by the input voltage Vin through the series circuit with the switch  306 , the inductive storage element  305  and the capacitive storage element  307 . When the switch  306  is switched off, the rectifier circuit  10  acts as a freewheeling element and enables the current I 1 , driven by the inductive storage element  305 , further to flow. 
     The rectifier circuit  10  may be operable to receive an external drive signal Sin. According to one embodiment, this drive signal Sin is provided by the control circuit  308 . In this case, the control circuit  308  is configured such that the switch  306  and the rectifier circuit  10  are not driven in the on-state at the same time. According to one embodiment, the control circuit  308  switches on the transistor in the rectifier circuit  10  each time the switch  306  is switched off. Further, the control circuit  308  is configured to switch off the transistor in the rectifier circuit  10  each time the switch  306  is switched off or each time the current I 1  decreases to zero. 
       FIG. 11  illustrates an embodiment of a power converter circuit including a flyback converter topology. Referring to  FIG. 11 , the power converter includes a transformer  405  with a primary winding  405   1  and a secondary winding  405   2 . The primary winding  405   1  is connected in series with a switch  406 , with the series circuit with the primary winding  405   1  and the switch  406  connected between input terminals  401 ,  402  for receiving an input voltage Vin. A series circuit with a rectifier circuit  10  and a capacitive storage element  407  is connected in parallel with the secondary winding  405   2 . An output voltage Vout is available across capacitive storage element  407  between output terminals  403 ,  404 . 
     Referring to  FIG. 11 , a control circuit  408  generates a drive signal S 406  of the switch  406  dependent on an output signal Sout. The output signal Sout is representative of the output voltage Vout. The drive signal S 406  is a pulse-width modulated (PWM) drive signal. The control circuit  408  adjusts the duty-cycle of the drive signal S 406  such that the output voltage Vout corresponds to a predefined set voltage. 
     The operating principle of the power converter circuit of  FIG. 11  is as follows: Each time the switch  406  is switched on, the primary winding  405   1  of the transformer  405  is connected between the input terminals  401 ,  402  and energy is magnetically stored in the primary winding  405   1 . A current I 1  through the secondary winding  405   2  is zero when the switch  406  is switched on, because the primary winding  405   1  and the secondary winding  405   2  have opposite winding senses. When the switch  406  is switched of, the primary winding transfers the energy previously stored therein to the secondary winding  405   2 , where a current I 1  through the secondary winding  405   2  flows through the rectifier circuit  10  to the output terminals  403 ,  404  and the capacitive storage element  407 , respectively. 
     The rectifier circuit  10  may be implemented in accordance with one of the embodiments explained before. The rectifier circuit  10  may be configured to receive an external drive signal Sin. This external drive signal Sin may be generated by the control circuit  408 . According to one embodiment, the drive signal Sin is generated such that the transistor in the rectifier circuit  10  is switched on when the switch  406  is switched off. Further, the external drive signal Sin may be generated such that the transistor in the rectifier circuit  10  is switched off, when the current I 1  decreases to zero or when the switch  406  is again switched on. 
       FIG. 12  illustrates a further embodiment of a power converter circuit. The power converter circuit of  FIG. 12  has a two transistor forward (TTF) topology. Referring to  FIG. 12 , the power converter includes a transformer  505  with a primary winding  505   1  and a secondary winding  505   2  that have identical winding senses. The primary winding  505   1  is connected between a first switch  506   1  and a second switch  506   2 , with the series circuit with the switches  506   1 ,  506   2  and the primary winding  505   1  connected between input terminals  501 ,  502  for receiving an input voltage Vin. A circuit node common to the first switch  506   1  and the primary winding  505   1  is coupled to the second input terminal  502  via a first rectifier element  507   1 , such as a diode. Further, a circuit node common to the primary winding  505   1  and the second switch  506   2  is coupled to the first input terminal  501  through a further rectifier element  507   2 , such as a diode. A series circuit with a first rectifier circuit  10   1 , an inductive storage element  508 , and a capacitive storage element  509  is connected in parallel with the secondary winding  505   2 . An output voltage Vout is available between output terminals  503 ,  504  across the capacitive storage element  509 . A further rectifier circuit  10   2  is connected in parallel with the series circuit with inductive storage element  508  and a capacitive storage  509 . 
     Referring to  FIG. 12 , a control circuit  510  generates a drive signal S 506  to the first and second switches  506   1 ,  506   2  that are synchronously switched on and switched off. The drive signal S 506  is a pulse-width modulated (PWM) drive signal that is dependent on an output signal Sout. This output signal Sout represents the output voltage Vout. The control circuit  510  generates the drive signal S 506  with a duty cycle such that the output voltage Vout corresponds to a predefined set voltage. 
     The operating principle of the power converter circuit of  FIG. 12  is as follows: Each time the first and second switches  506   1 ,  506   2  are switched on, the primary winding  505   1  is connected between the input terminals  501 ,  502  and a current I 505   1  flows through the primary winding. The polarity of a voltage V 505   2  across the secondary winding  505   2  is as indicated in  FIG. 12 . This voltage causes a current I 1   1  through the first rectifier circuit  10   1 , the inductive storage element  508  and the capacitive storage element  509 . When the switches  506   1 ,  506   2  are switched off, the current I 505   1  through the primary winding continuous to flow by virtue of the two rectifier elements  507   1 ,  507   2 . However, the polarity of the voltage V 505   2  is inverted, so that the current I 1   1  through the first rectifier circuit  10   1  becomes zero and a current I 1   2  through the second rectifier circuit  10   2  flows. 
     The first and second rectifier circuits  10   1 ,  10   2  may be implemented in accordance with one of the embodiments explained before. In particular, the rectifier circuits  10   1 ,  10   2  may be implemented to each receive an external drive signal Sin 1 , Sin 2  (illustrated in dashed lines in  FIG. 12 ), or may be configured to internally generate the drive signals. 
       FIG. 13  illustrates a further embodiment of a power converter circuit. The power converter circuit of  FIG. 13  includes a phase-shift zero-voltage switching (ZVS) full bridge topology. Referring to  FIG. 13 , the power converter circuit includes two half bridges each including a high-side switch  605   1 ,  606   1  and a low-side switch  605   2 ,  606   2  connected between input terminal  601 ,  602  for receiving an input voltage Vin. A series circuit with an inductive storage element  610  and a primary winding  607   1  of a transformer  607  is connected between output terminals of the two half bridges. The transformer  607  includes two secondary windings, namely a first secondary winding  607   2 , and a second secondary winding  607   3  that are inductively coupled with the primary winding  607   1 . The primary winding  607   1  and the secondary winding  607   2 ,  607   3  have identical winding senses. On the secondary side (the side with the secondary windings), the power converter circuit includes a series circuit with an inductive storage element  611  and a capacitive storage element  608 . The first primary winding  607   2  is coupled to this series circuit  611 ,  608 , through a first rectifier circuit  10   1 , and the second secondary winding  607   3  is coupled to the series circuit  611 ,  608  through a second rectifier circuit  10   2 . A third rectifier circuit  10   3  is connected in parallel with the series circuit with the inductive storage element  611  and the capacitive storage element  608 . Specifically, the inductive storage element  611  is connected to the first primary winding  607   2  through the first rectifier circuit  10   1  and to the second primary winding  607   3  through the second rectifier circuit  10   2 . A circuit node common to the first and second secondary winding  607   2 ,  607   3  is connected to that circuit node of the capacitive storage element  608  facing away from the inductive storage element  611  and to the second output terminal  604 , respectively. 
     The switches of the half-bridges are cyclically switched on and off by a drive circuit  609  dependent on an output signal Sout representing the output voltage Vout in accordance with a specific drive scheme. In  FIG. 13 , reference characters S 605   1 , S 605   2 , S 606   1 , S 606   2  denote drive signals provided by the drive circuit  609  to the individual switches  605   1 ,  605   2 ,  606   1 ,  606   2 . Each cycle in accordance with this drive scheme includes four different phases. In a first phase, the high-side switch  605   1  of the first half-bridge and the low-side switch  606   2  of the second half-bridge are switched on. Thus, a current I 607   1  flows through the first inductive storage element  610  and the primary winding  607   1 . Voltages V 607   2 , V 607   3  across the secondary windings  607   2 ,  607   3  have polarities as indicated in  FIG. 13 . The voltage V 607   2  causes a current I 1   1  through the first rectifier circuit  10   1 , the second inductive storage element  611  and the capacitive storage element  608 , while the second rectifier circuit  10   2  blocks. 
     In a second phase, the high side switch  605   1  of the first half-bridge is switched on and the high-side switch  606   1  of the second half-bridge is switched on. There may be a delay time between switching off the low-side switch  605   2  of the first half-bridge and switching on the high-side switch  606   1  of the second half-bridge. During this delay time, a freewheeling element (not illustrated) connected in parallel with the high-side switch  606   1  may take the current. The switches  605   1 ,  605   2 ,  606   1 ,  606   2  may be implemented as power MOSFETs, in particular as power MOSFETs that include an integrated body diode that may act as freewheeling element. 
     In the second phase, the voltage across the primary winding  607   1  and the voltages V 607   2 , V 607   3  across the secondary windings are zero. The current through the inductive storage element  611  continuous to flow, where the third rectifier circuit  10   3  takes the current through the inductive storage element  611  and the capacitive storage element  608 . 
     In the third phase, the high-side switch  606   1  of the second half-bridge and the low-side switch  605   2  of the first half-bridge are switched on. The voltages V 607   2 , V 607   3  across the secondary windings  607   2 ,  607   3  have polarities opposite to the polarities indicated in  FIG. 13 . In this case, a current flows through the second secondary winding  607   3 , the second rectifier circuit  10   2 , the inductive storage element  611  and the capacitive storage element  608 . 
     In the fourth phase, the low-side switch  605   2  of the first half-bridge is switched off, and the half-side switch  605   1  of the first half-bridge is switched on. The voltage across the primary winding  607   1  and the voltage across the secondary windings  607   2 ,  607   3  turn to zero. The current through the second inductive storage element  611  and the capacitive storage element  608  continuous to flow, where the third rectifier circuit  10   3  provides a current path for this current. 
     According to one embodiment, a timing of switching on and switching off the individual switches of the two half-bridges is such that at least some of the switches are switched on and/or switched off when the voltage across the respective switch is zero. 
     Each of the rectifier circuits  10   1 ,  10   2 ,  10   3  may be implemented in accordance with one of the embodiments explained before. In  FIG. 13 , reference characters  12   1 ,  12   2 ,  12   3  denote first load terminals and reference characters  13   1 ,  13   2 ,  13   3  denote second load terminals of the individual rectifier circuits  10   1 ,  10   2 ,  10   3 . 
       FIG. 14  illustrates a further embodiment of a power converter circuit. The power converter circuit of  FIG. 14  is implemented with a hard-switching half-bridge topology. This power converter circuit includes a half-bridge with a high-side switch  705   1  and a low-side switch  705   2  connected between input terminals  701 ,  702  for receiving an input voltage Vin. A capacitive voltage divider  706   1 ,  706   2  is also connected between the input terminals  701 ,  702 . A primary winding  707   1  of a transformer  707  is connected between an output terminal of the half-bridge and a center tap of the capacitive voltage divider. A secondary winding  707   2  of the transformer  707  and the primary winding  707   1  have same winding senses. A first terminal of the secondary winding  707   2  is connected to a first output terminal  703  through a first inductive storage element  708 , and a second terminal of the secondary winding  707   2  is connected to the first output terminal  703  through a second inductive storage element  709 . A capacitive storage element  710  is connected between the first output terminal  703  and a second output terminal  704 , where an output voltage Vout is available between these output terminals  703 ,  704 . The second output terminal  704  is connected to the first terminal of the secondary winding  707   2  through a first rectifier circuit  10   1 , and the second output terminal  704  is connected to the second terminal of the secondary winding  707   2  through a second rectifier circuit  10   2 . The first rectifier circuit  10   1  provides a freewheeling path for a first series circuit with the first inductive storage element  708  and the capacitive storage element  710 , and the second rectifier circuit  10   2  provides a freewheeling path for a second series circuit with the second inductive storage element  709  and the capacitive storage element  710 . 
     Each of the first and second rectifier circuit  10   1 ,  10   2  may be implemented in accordance with one of the embodiments explained herein before. In  FIG. 14 , reference characters  12   1 ,  12   2  denote first load terminals and reference characters  13   1 ,  13   2  denote second load terminals of the individual rectifier circuits  10   1 ,  10   2 . 
     A drive circuit  610  provides drive signals S 705   1 , S 705   2  for the switches  705   1 ,  705   2  of the half-bridge dependent on an output signal Sout. The output signal Sout represents the output voltage Vout. The drive signals S 705   1 , S 705   2  are generated such that the output voltage Vout corresponds to a predefined set value. 
     The operating principle of the power converter circuit of  FIG. 14  is as follows: The electrical potential at the center tap of the capacitive voltage divider  706   1 ,  706   2  is somewhere between electrical potentials at the first and second input terminals  701 ,  702 . Just for explanation purposes it is assumed that the electrical potential at the center tap corresponds to half the input voltage Vin. 
     Each time the high-side switch  705   1  of the half-bridge is switched on, a voltage across the primary winding  707   1  is positive and a resulting voltage V 707   2  across the secondary winding  707   2  has the polarity as indicated in  FIG. 14 . In this case, a current flows through the first inductive storage element  708 , the capacitive storage element  707 , the second rectifier circuit  10   2  and the secondary winding  707   2 . During this phase, energy is magnetically stored in the first inductive storage element  708 . 
     In a second phase, both switches are switched off. In this phase, the current through the first inductive  708  continuous to flow, where the first rectifier circuit  10   1  connected between the second output terminal  704  and the first inductive storage element  708  takes the current. 
     In a third phase, low side switch  705   2  of the half-bridge is switched on. A voltage across the primary winding  707   1  is negative in this case, and the corresponding voltage V 707   2  across the secondary winding  707   2  has a polarity opposite to the polarity indicated in  FIG. 14 . In this case, the current flows through the secondary winding  707   2 , the second inductive storage element  709 , the output capacitance  710  and the first rectifier circuit  10   1 . 
     In a fourth phase, both switches  705   1 ,  705   2  are switched off. In this phase, the current through the second inductive storage element  709  continuous to flow, where the second rectifier circuit  10   2  takes the current in this case. 
       FIG. 15  illustrates a power converter circuit according to a further embodiment. The power converter circuit of  FIG. 15  includes an LLC resonant topology. Referring to  FIG. 15 , the power converter circuit includes a half-bridge with a high-side switch  805   1  and a low-side switch  805   2  connected between the input terminals  801 ,  802  for receiving an input voltage Vin. The power converter circuit further includes a series LLC circuit with a capacitive storage element  806 , an inductive storage element  807 , and a primary winding  809   1  of a transformer  809  connected in parallel with the low-side switch  805   2 . A further inductive storage element  808  is connected in parallel with the primary winding  809   1 . The transformer  809  includes two primary secondary windings, namely a first secondary winding  809   2  and a second secondary winding  809   3  coupled to the primary winding  809   1  and each having same winding sense as the primary winding  809   1 . The first secondary winding  809   2  is coupled to a first output terminal  803  through a first rectifier circuit  10   1 , and the second primary winding  809   3  is coupled to the first output terminal  803  through the second rectifier circuit  10   2 . A circuit node common to the first and second secondary windings  809   2 ,  809   3  is coupled to a second output terminal  804 . A capacitive storage element  810  is connected between the output terminals  803 ,  804 , where an output voltage Vout is available between the output terminals  803 ,  804 . 
     In  FIG. 15 , S 805   1 , S 805   2  denotes drive signals for the switches  805   1 ,  805   2  of the half-bridge. These drive signals S 805   1 , S 805   2  are generated by a drive circuit  811  in accordance with an output signal Sout. The output signal Sout represents the output voltage Vout. The drive circuit  8  is configured to generate the drive signals S 805   1 , S 805   2  such that the output voltage Vout corresponds to a predefined set value. 
     In the power converter circuit of  FIG. 15 , the high-side switch  805   1  and the low-side switch  805   2  are switched on and off alternatingly. This causes an alternating current through the primary winding  809   1  of the transformer  809 . This alternating current is transferred to the secondary side. When the alternating current through the primary winding  809   1  has a first direction, a current on the secondary side flows through the first primary winding  809   2  and the first rectifier circuit  10   1  to the capacitive storage element  810  and the output terminals  803 ,  804  respectively. When the current through the primary winding  809   1 , has an opposite second direction, the current on the secondary side flows through the second secondary winding  809   3  and the second rectifier circuit  10   2  to the capacitive storage element  810  and the output terminals  803 ,  804 , respectively. 
     In  FIG. 15 , reference characters  12   1 ,  12   2  denote first load terminals of the first and second rectifier circuits  10   1 ,  10 , and reference characters  13   1 ,  13   2  denote second load terminals of the first and second rectifier circuits  10   1 ,  10   2 . Each of these rectifier circuits  10   1 ,  10   2  may be implemented in accordance with one of the embodiments explained herein before. 
     In each of the power converter circuits explained before, a load (not illustrated) may be connected to the output terminals to receive the output voltage Vout. 
     In case one of the power converter circuits explained before, includes more than one rectifier circuit, the individual rectifier circuits may be implemented identically. However, it is also possible to implement two or more rectifier circuits in one power converter circuit with different topologies. 
       FIG. 16  illustrates a further embodiment of a circuit arrangement including a rectifier circuit  10 . The circuit arrangement includes input terminals  901 ,  902  for receiving an input voltage Vin, a series circuit with a load Z and a switch  903  connected between the input terminals  901 ,  902  and a rectifier circuit  10  connected in parallel with the load Z. The load Z is, e.g., an inductive load. That is, the load Z includes at least one inductive element or a circuit element with an inductive behavior. The switch  903  is a low-side switch. That is, the switch  903  is connected between the load Z and the terminal for the negative supply potential or reference potential of the input voltage Vin. A circuit configuration as illustrated in  FIG. 16  may, e.g. be implemented in a current controller for controlling a current through an inductive load. 
     The operating principle of the circuit arrangement of  FIG. 16  is as follows: Each time the switch  903  is switched on, the load Z is connected between the input terminals  901 ,  902  and a current I 1  flows through the load Z. When the switch  903  is switched off, the current I 1  through the load Z by virtue of the inductive character of the load continues to flow (and decreases). In this phase, the rectifier circuit  10  acts as a freewheeling element and takes the current I 1  flowing through the load Z. 
     The switch  903  is switched on and off by a drive signal S 903  provided by a control circuit  904 . According to one embodiment, the control circuit  904  is configured to adjust a duty-cycle of the drive signal S 903  dependent on the voltage I 1  through the load Z in order to control an average value of the current I 1  through the load to correspond to a predefined set value. 
       FIG. 17  illustrates the circuit arrangement of  FIG. 16  that includes a rectifier circuit in accordance with the embodiment of  FIG. 5 . The switch  903  is implemented similar to the rectifier element  1  of the rectifier circuit  10  with a first transistor  2   903  and with an arrangement  30   903  with a plurality of second transistors. In the embodiment of  FIG. 17 , first transistor  20   903  of the switch  903  is implemented as an n-type enhancement MOSFET. However, this is only an example. The switch  903  could be implemented with any other type of first transistor as well. The arrangement  30   903  with the second transistors may be implemented like the arrangement  30  with the second transistors  3   1 - 3   n  explained in connection with the rectifier element  1  in  FIG. 2  before. The operating principle of the switch  903  corresponds to the operating principle of the rectifier element of  FIG. 2 . That is, the switch  1  is in the on-state (switched on) when the first transistor  2   903  is switched on, and the switch  903  is in the off-state (switched off) when the first transistor  2   903  is switched off. The drive signal S 903  received from the control circuit (not illustrated in  FIG. 17 ) is configured to one of switch on and switch off the first transistor  2   903 . 
       FIG. 18A  illustrates one embodiment of a detection circuit  9  of the rectifier circuit  10  in the circuit arrangement of  FIG. 17 . In  FIG. 18 , only some of the circuit elements of the rectifier element  1  of the rectifier circuit  10  and only some of the circuit elements of the switch  903  are illustrated, namely those circuit elements necessary for understanding the operating principle of the detection circuit  9 .  FIG. 18  shows the first transistor  2 , the body diode D 2  and the optional voltage limiting element  7   0  of the rectifier element  1  and an n-th second transistor  3   n−903  of the switch  903 . The function of this second transistor  3   n−903  corresponds to the function of the second transistor  3   n  illustrated in  FIG. 2 . Reference character  7   n−903  denotes the optional voltage limiting element connected in parallel with this second transistor  3   n−903 . 
     Referring to  FIG. 18A , the detection circuit  9  includes an amplifier  92   4 , such as an operational amplifier (OA). This amplifier  92   4  is operable to evaluate a voltage across the body diode D 2  of the first transistor  2  of the rectifier element  1  in order to determine a current I 1  through the rectifier element  1 . A first load terminal  22  (corresponding to the anode terminal of the body diode D 2 ) of the first transistor  2  is coupled to a first terminal of the operational amplifier  92   4  through a first resistive element  92   1 , and the second load terminal  23  of the second transistor  2  is coupled to the first terminal of the amplifier  92   4  through a second resistive element  92   2 . Further, that load terminal of the second transistor  3   n−903  facing away from the first transistor  2  is coupled to a second terminal of the amplifier  92   4  through a third resistive element  92   3 . The second terminal of the amplifier  92   4  is coupled to the output terminal through a further resistive element  92   5 . The detection signal S D  is available at the output of the amplifier  92   4 . Optionally, buffers  92   6 ,  92   7 ,  92   8  are connected between the first, second and third resistive elements and the corresponding circuit nodes of the rectifier element  1  and the switch  903 . The output signal S D  of the amplifier  92   4  represents the direction of the current I 1 , where the output signal S D  has a first sign when the current flows in the first direction and has a second sign when the current flows in the opposite second direction. 
       FIG. 18B  illustrates a modification of the detection circuit  9  of  FIG. 18A . The detection circuit  9  of  FIG. 18B  includes two shunt resistors, a first shunt resistor  92   9  between the first load terminal  12  of the rectifier circuit  10  and the circuit node for connecting the load Z thereto, and a second shunt resistor  92   9  between the circuit node for connecting the load Z thereto and the switch  903 . In this detection circuit  9 , the first input terminal (the non-inverting terminal) of the amplifier  92   4  is coupled to the circuit node common to the first shunt resistor  92   9  and the rectifier circuit via the second resistor  92   9  and to the circuit node common to the first shunt resistor  92   9  and the second shunt resistor  92   10  via the first resistor  92   10 . Like in the embodiment of  FIG. 18A , the buffers  92   6 ,  92   7  are optional. The second input terminal (the inverting terminal) of the amplifier  92   4  is coupled to the circuit node common to the second shunt resistor  92   10  and the switch  903 . In this detection circuit  9 , the detection signal S D  at the output of the amplifier  92   4  represents the direction of the current I 1  through the rectifier circuit  10  and the amplitude of the current I 1 . 
       FIG. 19  illustrates a further embodiment of a detection circuit  9 . The detection circuit  9  of  FIG. 19  is based on the detection circuit  9  of  FIG. 18B  and further includes a differentiator  93  receiving the current measurement signal at the output of the amplifier  92   4 . In  FIG. 19 , reference character S 92   4  denotes the output signal of the amplifier that corresponds to the detection signal of  FIG. 18 . The differentiator  93  may be implemented like a conventional differentiator. Just for illustration purposes one embodiment of the differentiator  93  is illustrated in detail in  FIG. 19 . 
     The differentiator  93  of  FIG. 19  includes a further amplifier  93   1 , such as an operational amplifier (OA). The output of the amplifier  92   4  is coupled to a first input (the inverting input in this embodiment) of the further amplifier  93   1  through a capacitive element  93   2 . Further, the inverting input is coupled to the output through a resistor  93   3 . A detection signal S D  at the output of the differentiator  93  corresponds to a voltage between the output of the further amplifier  93   1  and the second input terminal (the non-inverting input terminal in this embodiment) of the further amplifier  93   1 . This output signal corresponds to a time derivative of the current measurement signal S 92   4  at the output of the amplifier  92   4 . The time derivative of the current measurement signal S 92   4  is positive when the current I 1  through the rectifier circuit  1  increases, and is negative when the current through the rectifier circuit decreases. 
     The control and drive circuit  8  (not illustrated in  FIG. 19 ) receiving the detection signal S D  of  FIG. 19  may be configured to detect maxima of the detection signal S D  and may be configured to switch on the first transistor  2  of the rectifier circuit  1  when the detection signal S D  has a positive maximum, and may be configured to switch off the first transistor  2  of the rectifier circuit  1  when the detection signal S D  has a negative maximum. 
     Optionally, a rectifier  94  is connected downstream the output the further amplifier  93   1 . The rectifier  94  receives the detection signal S D  and provides a rectified detection signal |S D |. 
       FIG. 20  illustrate a modification of the circuit arrangement of  FIG. 17 . In the circuit arrangement of  FIG. 20 , the rectifier circuit  10  is operable to receive an external drive circuit Sin. This external drive signal Sin is provided from a control circuit  94  through a level shifter  95 . The control circuit  94  may also provide the drive signal of the switch  903 . The level shifter  95  includes a series circuit with a first transistor  2   95  receiving the drive signal S in  and a plurality of n (with n&gt;1) second transistors  3   1−95 - 3   n−95  connected in series with the first transistor. The series circuit with the first transistor  2   95  and the second transistors  3   1−95 - 3   n−95  is connected between the terminal  902  for the reference potential and the circuit node between the first transistor  2  of the rectifier circuit  10  and the arrangement  30  with the second transistors. Referring to  FIG. 20 , the first transistor  2   95  of the level shifter may be implemented as an enhancement MOSFET, specifically an n-type enhancement MOSFET, while the second transistors  3   1−95 - 3   n−95  may be implemented as depletion MOSFETs (or JFETs). Each of the second transistors  3   1−95 - 3   n−95  has its gate terminal connected to its source terminal, wherein the source terminal of the 1st second transistor  3   1−95  is connected to the drain terminal of the first transistor. A voltage limiting element  7   0−95 - 7   n−95 , such as a Zener diode or a series circuit of Zener diodes, is connected in parallel with the first transistor  2   95  and each of the second transistors  3   1−95 - 3   n−95 . 
     An evaluation circuit  95   1 - 95   3  compares the electrical potential at the load terminal of one of the second transistors, namely the upper second transistor  7   n−95  in this embodiment, with the electrical potential at the first load terminal of the rectifier circuit  10  and generates the drive signal S 2  for the first transistor  2  of the rectifier circuit  10  dependent on the comparison. The electrical potential at the second transistor  7   n−95  is dependent on the switching state of the first transistor  2   95  of the level shifter  95 . This electrical potential is a high electrical potential when the first transistor  2   95  is switched on and is a low electrical potential when the first transistor  2   95  is switched on. Thus, by switching on and switching off the first transistor  2   95  different electrical potentials are generated at the second transistor  3   n−95  where this electrical potential is used to generate the drive signal of the first transistor  2  in the rectifier circuit  10 . Referring to  FIG. 20 , the evaluation circuit includes an amplifier  95   1  with a first (non-inverting) input coupled to the first load terminal  12  of the rectifier circuit  10 , and with a second (inverting) input coupled to the load terminal (source terminal) of the second transistor  3   n−95  through a resistor  95   2  and coupled to the output through a further resistor  95   3 . The drive signal S 2  is available at the output of the amplifier  95   1 . 
     The first semiconductor device  2  and the second semiconductor devices (second transistors)  3  that are represented by circuit symbols in the figures explained above can be implemented in many different ways. Some illustrative embodiments for implementing the second transistors  3  are explained with reference to Figures below. 
       FIG. 21A  shows a perspective view of one second transistor  3 .  FIG. 21B  shows a vertical cross sectional view and  FIG. 21C  shows a horizontal cross sectional view of this second transistor  3 .  FIGS. 21A ,  21 B,  21 C only show that section of the semiconductor body  100  in which the second transistor  3  is implemented. Active regions of the first semiconductor device  2  and active regions of neighbouring second transistors are not shown. The second transistor  3  according to  FIGS. 21A to 21C  is implemented as a MOSFET, specifically as a FINFET, and includes a source region  53 , a drain region  54  and a body region  55  that are each arranged in a fin-like semiconductor section  52 , which will also be referred to as “semiconductor fin” in the following. The semiconductor fin is arranged on a substrate  51 . In a first horizontal direction, the source and drain regions  53 ,  54  extend from a first sidewall  52   2  to a second sidewall  52   3  of the semiconductor fin  52 . In a second direction perpendicular to the first direction the source and drain regions  53 ,  54  are distant from one another and are separated by the body region  55 . The gate electrode  56  (illustrated in dashed lines in  FIG. 21A ) is dielectrically insulated from the semiconductor fin  52  by a gate dielectric  57  and is adjacent to the body region  55  on the sidewalls  52   2 ,  52   3  and on a top surface  52   1  of semiconductor fin  52 . 
       FIGS. 22A to 22C  illustrate a further embodiment of one second transistor  3  implemented as a FINFET.  FIG. 22A  shows a perspective view,  FIG. 22B  shows a vertical cross sectional view in a vertical section plane E-E, and  FIG. 22C  shows a horizontal cross sectional view in horizontal section plane D-D. The vertical section plane E-E extends perpendicular to the top surface  52   1  of the semiconductor fin  52  and in a longitudinal direction of the semiconductor fin  52 . The horizontal section plane D-D extends parallel to the top surface  52   1  of the semiconductor fin. The “longitudinal direction” of the semiconductor fin  52  corresponds to the second horizontal direction and is the direction in which the source and drain region  53 ,  54  are distant from one another. 
     The transistor  3  according to  FIGS. 22A to 22C  is implemented as a U-shape-surround-gate-FINFET. In this transistor, the source region  53  and the drain region  54  extend from the first sidewall  52   2  to the second sidewall  52   3  of the semiconductor fin  52  in the first horizontal direction, and are distant from one another in the second horizontal direction (the longitudinal direction of the semiconductor fin  52 ) that is perpendicular to the first horizontal direction. Referring to  FIGS. 22A and 22B , the source region  53  and the drain region  54  are separated by a trench which extends into the body region  55  from the top surface  52   1  of the semiconductor fin and which extends from sidewall  52   2  to sidewall  52   3  in the first horizontal direction. The body region  55  is arranged below the source region  53 , the drain region  54  and the trench in the semiconductor fin  52 . The gate electrode  56  is adjacent to the body region  55  in the trench and along the sidewalls  52   2 ,  52   3  of the semiconductor fin  52  and is dielectrically insulated from the body region  55  and the source and drain regions  53 ,  54  by the gate dielectric  57 . In an upper region of the trench, which is a region in which the gate electrode  56  is not arranged adjacent to the body region  55 , the gate electrode  56  can be covered with an insulating or dielectric material  58 . 
     The second transistors of  FIGS. 21A to 21C  and of  FIGS. 22A to 22C  are, for example, implemented as depletion transistors, such as an n-type or a p-type depletion transistors. In this case, the source and drain regions  53 ,  54  and the body region  55  have the same doping type. The body region  55  usually has a lower doping concentration than the source and drain regions  53 ,  54 . The doping concentration of the body region  55  is, e.g., about 2E18 cm −3 . In order to be able to completely interrupt a conducting channel in the body region  55  between the source region  53  and the drain region  54 , the gate electrode  56  along the sidewalls  52   2 ,  52   3  of the semiconductor fin  52  completely extends along the semiconductor fin  52  in the second horizontal direction (the longitudinal direction). In the vertical direction the gate electrode  56  along the sidewalls  52   2 ,  52   3  extends from the source and drain regions  53 ,  54  to at least below the trench. 
     Referring to  FIGS. 21A and 22A , the source region  53  is connected to the first load terminal (source terminal)  32 , the drain region  54  is connected to the second load terminal (drain terminal)  33 , and the gate electrode  56  is connected to the control terminal (gate terminal)  31 . These terminals are only schematically illustrated in  FIGS. 21A and 22A . 
     A thickness of the semiconductor fin  52 , which is the dimension of the semiconductor fin in the first horizontal direction, and the doping concentration of the body region  55  are adjusted such that a depletion region controlled by the gate electrode  56  can extend from sidewall  52   2  to sidewall  52   3  in order to completely interrupt a conducting channel between the source and the drain region  53 ,  54  and to switch the second transistor  3  off. In an n-type depletion MOSFET a depletion region expands in the body region  55  when a negative control (drive) voltage is applied between the gate electrode  56  and the source region  53  or between the gate terminal  31  and the source terminal  32 , respectively. Referring to the explanation provided with reference to  FIG. 1 , this drive voltage is dependent on the load voltage of the first semiconductor device  2 , or is dependent on the load voltage of another one of the second transistors  3 . How far the depletion region expands perpendicular to the sidewalls  52   2 ,  52   3  is also dependent on the magnitude of the control voltage applied between the gate terminal  31  and the source terminal  32 . Thus, the thickness of the semiconductor fin  52  and the doping concentration of the body region  55  are also designed dependent on the magnitude of the control voltage that can occur during the operation of the semiconductor device arrangement. 
     Implementing the FINFETs illustrated in  FIGS. 21A to 21C  and  22 A to  22 C as U-shape-surround-gate-FINFET, in which the channel (body region)  55  has an U-shape and the gate electrode  56  is also arranged on sidewalls  52   2 ,  52   3  and on a top surface  52   1  of the semiconductor fin  130  is only an example. These FINFETs could also be modified (not illustrated) to have the gate electrode  56  implemented with two gate electrode sections arranged on the sidewalls  52   2 ,  52   3  but not on the top surface  52   1  of the semiconductor fin  52 . A FINFET of this type can be referred to as double-gate FINFET. Each of the FINFETs explained above and below can be implemented as U-shape-surround-gate-FINFET or as double-gate FINFET. It is even possible to implement the individual second transistors  3  as different types of MOSFETs or FINFETs in one integrated circuit. 
     Each of the second transistors  3  and the first semiconductor device  2  can be implemented as FINFET. These individual FINFETs can be implemented in different ways to form the semiconductor arrangement  1 . 
       FIG. 23  illustrates a vertical cross sectional view of a semiconductor fin  52  in which active regions (source, drain and body regions) of a first semiconductor device  2  and of n second transistors  3  are arranged. In this embodiment, the first semiconductor device  2  and the second transistors are implemented as U-shape-surround-gate FINFET or as double-gate FINFET. In  FIG. 23 , like reference numbers are used to denote like features as in  FIGS. 21A to 21C  and  22 A to  22 C. In  FIG. 23  the reference numbers of like features of the different second transistors  3   1 - 3   n  have different indices (1, 2, 3, n). 
     Referring to  FIG. 23 , the active regions of neighboring second transistors  3  are insulated from each other by dielectric layers  59  which extend in a vertical direction of the semiconductor fin  52 . These dielectric layers  59  may extend down to or down into the substrate  51 . Further, the dielectric layers  59  extend from sidewall to sidewall of the semiconductor fin  52 . However, this is out of view in  FIG. 23 . The active regions of the first semiconductor device  2  are dielectrically insulated from active regions of the 1st second transistor  3   1  by a further dielectric layer  66  that also extends in a vertical direction of the semiconductor fin  52 . In the first semiconductor device  2 , a source region  61  and a drain region  62  are separated by a body region  63 . The gate electrode  64  that is arranged in the trench (and the position of which at the sidewalls of the semiconductor fin is illustrated by dotted lines), extends from the source region  61  along the body region  63  to the drain region  62 . The source region  61  is connected the first load terminal  22  that forms the first load terminal  12  of the semiconductor arrangement  1 , the drain region  62  is connected to the second load terminal  23 , and the gate electrode  64  is connected to the control terminal  21  that forms the control terminal  11  of the semiconductor arrangement  1 . The body region  63  is also connected to the first load terminal  22 . 
     The first semiconductor device  2  is, for example, implemented as an enhancement MOSFET. In this case, the body region  63  is doped complementarily to the source and drain regions  61 ,  62 . In an n-type MOSFET, the source and drain regions  61 ,  62  are n-doped while the body region  63  is p-doped, and in a p-type MOSFET, the source and drain regions  61 ,  62  are p-doped while the body region  63  is n-doped. 
     According to one embodiment, the substrate  51  is doped complementarily to the active regions of the second transistors  3  and to the source and drain regions  61 ,  62  of the first semiconductor device  2 . In this case, there is a junction isolation between the individual second transistors  3 . According to a further embodiment (illustrated in dashed lines), the substrate is an SOI substrate and includes a semiconductor substrate  51   1  and an insulation layer  51   2  on the semiconductor substrate  51   1 . The semiconductor fin  52  is arranged on the insulation layer. In this embodiment, there is a dielectric layer between the individual second transistors  3  in the substrate  51 . 
     According to yet another embodiment, illustrated in  FIG. 24 , the substrate  51  has the same doping type as the active regions of the second transistors  3  and as the source and drain regions  61 ,  62  of the first semiconductor device  2 . In this embodiment, the gate electrode  56  of the first semiconductor device  2  extends to the substrate, so that there is a conducting path in the body region between the source region  61  and the substrate  51  when the first semiconductor device  2  is in the on-state. Further the substrate is connected to the second load terminal  13  of the semiconductor arrangement through a contact region  67  of the same doping type as the substrate  51 . The contact region  67  is more highly doped than the substrate  51  and extends from the first surface  52   1  of the semiconductor fin  52  to the substrate. The contact region  67  may adjoin the drain region  54   n  of the n-th second transistor  3 . The contact region  67  is optional. A connection between the second load terminal  13  and the substrate  51  could also be provided through the drain and body regions  54   n ,  55   n  of the second transistor  3   n . 
     In the semiconductor arrangement of  FIG. 24 , the substrate  51  forms a current path that is parallel to the current path through the second transistors  3  or that is parallel to the ADZ. The substrate  51  is similar to the drift region in a conventional power transistor. In this embodiment, the body regions  55  of the individual second transistors  3  are coupled to the drift region  51 . 
     According to further embodiment (illustrated in dashed lines in  FIG. 24 ) the substrate  51  includes a semiconductor layer  51   3  doped complementary to remaining sections of the substrate  51  and to the body regions  55  of the second transistors  3 . This layer  51   3  is arranged between the body regions  55  of the second transistors  3  and those sections of the substrate acting as a drift region and provides a junction insulation between the individual second transistors  3  in the substrate  51 . 
     The semiconductor arrangement  1  of  FIG. 3  with the diode  2  connected in series with the second transistors  3  can easily be obtained from the arrangements illustrated in  FIGS. 21 and 22  by either connecting the control terminal of the first semiconductor device to the first load terminal  22  or by let the control terminal  21  floating. In this case, only the body diode of the MOSFET, which is the diode formed by the pn-junction between the body region  63  and the drain region  65  is active between the first and second load terminals  22 ,  23  of the second semiconductor device. 
     Each of the first semiconductor device  2  and the second transistors  3  (referred to as devices in the following) may include a plurality of identical cells (transistor cells) that are connected in parallel. Each of these cells can be implemented like the first semiconductor device  2  or like the second transistors  3 , respectively, illustrated in  FIGS. 21 and 22 . Providing a plurality of cells connected in parallel in one device can help to increase the current bearing capability and to reduce the on-resistance of the individual device. 
       FIG. 25  illustrates a top view on a semiconductor arrangement according to a first embodiment which includes a first semiconductor device  2  and a plurality of second transistors  3 , with each of these devices having a plurality (from which three are illustrated) cells connected in parallel. The individual cells of one device are implemented in different semiconductor fins  52   I ,  52   II ,  52   III . Each of these cells has a source region  61 ,  53  that is additionally labeled with “S” in  FIG. 25 , and a drain region  62 ,  54  that is additionally labeled with “D” in  FIG. 25 . The cells of one device are connected in parallel by having the source regions of the one device connected together and by having the drain regions of the one device connected together. These connections as well as connections between the load terminals of the different devices are schematically illustrated in bold lines in  FIG. 25 . Connections between the control terminals (gate terminals) and the load terminals of the different devices are not illustrated in  FIG. 25 . The connections between the cells and the different devices can be implemented using conventional wiring arrangements arranged above the semiconductor body and contacting the individual active regions (source and drain regions) through vias. Those wiring arrangements are commonly known so that no further explanations are required in this regard. The individual cells of one device  2 ,  3   1 ,  3   2 ,  3   3 ,  3   n  have a common gate electrode  64 ,  56   1 ,  56   2 ,  56   3 ,  56   n  arranged in the U-shaped trenches of the individual semiconductor fins and in trenches between the individual fins. These “trenches between the fins” are longitudinal trenches along the fins. All gates  64 ,  56   1 ,  56   2 ,  56   3 ,  56   n  are electrically isolated from each other by a dielectric  66  and  59 . 
       FIG. 26  illustrates a further embodiment for implementing one second transistor  3  with a plurality of transistor cells. In this embodiment, a plurality of transistor cells of the second transistor  3  are implemented in one semiconductor fin. In the longitudinal direction of the semiconductor fin  52 , source and drain regions  53 ,  54  are arranged alternatingly with a source region  53  and a neighboring drain region  54  being separated by one (U-shaped) trench that accommodates the gate electrode  56 . The source regions  53  are connected to the first load terminal  22 , and the drain regions  54  are connected to the second load terminal  23 , so that the individual transistor cells are connected in parallel. The gate electrode  56  is common to the individual transistor cells and extends along the sidewalls of the semiconductor fin  52  in the longitudinal direction. Each source region  53  and each drain region  54  (except for the source and drain regions arranged at the longitudinal ends of the semiconductor fin  52 ) is common to two neighboring transistor cells. 
     The concept of providing several transistor cells in one semiconductor fin explained with reference to  FIG. 26  is, of course, also applicable to the implementation of the first semiconductor device  2 . 
     Referring to  FIGS. 27A to 27C , one second transistor  3  may include a plurality of semiconductor fins  52   IV ,  52   V ,  52   VI ,  52   VII , with each semiconductor fin  52   IV - 52   VII  including a plurality of transistor cells (one of these cells is highlighted by a dashed and dotted frame in  FIG. 27A ).  FIG. 27A  shows a top view of one second transistor  3 ,  FIG. 27B  shows a vertical cross sectional view in a section plane F-F cutting through source regions in different fins, and  FIG. 27C  shows a vertical cross sectional view in a section plane G-G cutting through the trenches with the gate electrode  56  in different fins. Referring to  FIG. 27A , the source regions of the individual transistor cells are connected to the first load terminal  22  and the drain regions of the individual transistor cells are connected to the second load terminal  23  so that the individual transistor cells are connected in parallel. These connections are only schematically illustrated in  FIG. 27A . 
     The concept of providing a plurality of semiconductor fins with each semiconductor fin including a plurality of transistor cells explained with reference to  FIGS. 27A to 27C  is, of course, also applicable to the implementation of the first semiconductor device  2 . 
     Although only 20 transistor cells are illustrated in  FIG. 27A , namely five cells in each of the four semiconductor fins  52   IV - 52   VII , one second transistor  3  or the first semiconductor device  2  may include up to several thousand or even up to several ten or several hundred million transistor cells connected in parallel. The individual transistor cells form a matrix of transistor cells that are connected in parallel. A device (first semiconductor device  2  or second transistor  3 ) having a plurality of transistor cells arranged in a matrix will be referred to as matrix device in the following. 
       FIG. 28  illustrates how second transistors implemented as matrix devices can be connected in series. For illustration purposes, only two second transistors  3   i ,  3   i+1  are shown in  FIG. 28 . For connecting these two transistors in series, the source regions of the second transistor  3   i+1  are connected to the drain regions of the transistor  3   i . The source regions of the second transistor  3   i  are connected to the drain regions of second transistors  3   i−1  (not illustrated), and the drain regions of the second transistor  3   i+1  are connected to the source regions of second transistors  3   i+2  (not illustrated). 
       FIG. 29  illustrates a vertical cross sectional view of a transistor cell of the first transistor  2  according to a further embodiment. Several of the transistor cells of  FIG. 19  may be connected in parallel to form the first transistor  2 . The transistor cell of  FIG. 19  is implemented with a planar gate electrode  64 . The gate electrode  64  is arranged above the first surface  101  of the semiconductor body  100  and is dielectrically insulated from the body region  63  by the gate dielectric  65 . The source and drain regions  61 ,  62  are arranged in the region of the first surface  101  and are distant in a lateral direction of the semiconductor body  100 . The body region  63  adjoins the substrate  51 , where the substrate  51  may be implemented in accordance with one of the embodiments explained before. Further, the body region  63  is electrically connected to the source terminal  22 . Referring to  FIG. 19 , the vertical dielectric layer  66  may extend through the body region  63  to or into the substrate  51 . The vertical dielectric layer  66  may surround the body region  63  in a horizontal plane of the semiconductor body  100 , which is a plane perpendicular to the section plane illustrated in  FIG. 19 . The first transistor  2  of  FIG. 19  may be implemented as an enhancement transistor. In this case, the body region  63  is doped complementary to the source and drain regions  61 ,  62 . Concerning the doping types of the individual device regions reference is made to the embodiments explained before. 
       FIG. 30  illustrates a vertical cross sectional view of a transistor cell of one second transistor  2  according to a further embodiment. Several of the transistor cells of  FIG. 20  may be connected in parallel to form one second transistor  3 . The transistor cell of  FIG. 20  is implemented with a planar gate electrode  56 . The gate electrode  56  is arranged above the first surface  101  of the semiconductor body  100  and is dielectrically insulated from the body region  55  by the gate dielectric  57 . The source and drain regions  53 ,  54  are arranged in the region of the first surface  101  and are distant in a lateral direction of the semiconductor body  100 . The body region  55  adjoins the substrate  51 , where the substrate  51  may be implemented in accordance with one of the embodiments explained before. Further, the body region  55  is electrically connected to the source terminal  32 . Referring to  FIG. 20 , the vertical dielectric layer  59  may extend through the body region  55  to or into the substrate  51 . The vertical dielectric layer  59  may surround the body region  55  in a horizontal plane of the semiconductor body  100 , which is a plane perpendicular to the section plane illustrated in  FIG. 20 . 
     The second transistor  3  of  FIG. 20  may be implemented as a depletion transistor. In this case, the body region  55  is doped complementary to the source and drain regions  53 ,  54  and includes a channel region  55 ′ of the same doping type as the source and drain regions  53 ,  54  along the gate dielectric  57 . The channel region  55 ′ extends from the source region  53  to the drain region  54 . In an n-type depletion transistor, the source region  53 , the drain region  54  and the channel region  55 ′ are n-doped while the body region is p-doped. In a p-type depletion transistor, the doping types of these device regions are complementary to those in an n-type transistor. 
     Although various exemplary embodiments of the invention have been disclosed, it will be apparent to those skilled in the art that various changes and modifications can be made which will achieve some of the advantages of the invention without departing from the spirit and scope of the invention. It will be obvious to those reasonably skilled in the art that other components performing the same functions may be suitably substituted. It should be mentioned that features explained with reference to a specific figure may be combined with features of other figures, even in those cases in which this has not explicitly been mentioned. Further, the methods of the invention may be achieved in either all software implementations, using the appropriate processor instructions, or in hybrid implementations that utilize a combination of hardware logic and software logic to achieve the same results. Such modifications to the inventive concept are intended to be covered by the appended claims. 
     Spatially relative terms such as “under”, “below”, “lower”, “over”, “upper” and the like, are used for ease of description to explain the positioning of one element relative to a second element. These terms are intended to encompass different orientations of the device in addition to different orientations than those depicted in the figures. Further, terms such as “first”, “second”, and the like, are also used to describe various elements, regions, sections, etc. and are also not intended to be limiting. Like terms refer to like elements throughout the description. 
     As used herein, the terms “having”, “containing”, “including”, “comprising” and the like are open ended terms that indicate the presence of stated elements or features, but do not preclude additional elements or features. The articles “a”, “an” and “the” are intended to include the plural as well as the singular, unless the context clearly indicates otherwise. 
     It is to be understood that the features of the various embodiments described herein may be combined with each other, unless specifically noted otherwise. 
     Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.