Patent Publication Number: US-9847900-B2

Title: Receiver and method of receiving

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     The present application claims priority to United Kingdom Application 1512952.1 filed on 22 Jul. 2015, the contents of which being incorporated herein by reference in its entirety. 
     FIELD OF THE DISCLOSURE 
     The present disclosure relates to receivers and methods of receiving payload data using Orthogonal Frequency Division Multiplexed (OFDM) symbols. 
     BACKGROUND OF THE DISCLOSURE 
     There are many examples of radio communications systems in which data is communicated using Orthogonal Frequency Division Multiplexing (OFDM). Television systems which have been arranged to operate in accordance with Digital Video Broadcasting (DVB) standards for example, use OFDM for terrestrial and cable transmissions. OFDM can be generally described as providing K narrow band sub-carriers (where K is an integer) which are modulated in parallel, each sub-carrier communicating a modulated data symbol such as for example Quadrature Amplitude Modulated (QAM) symbol or Quaternary Phase-shift Keying (QPSK) symbol. The modulation of the sub-carriers is formed in the frequency domain and transformed into the time domain for transmission. Since the data symbols are communicated in parallel on the sub-carriers, the same modulated symbols may be communicated on each sub-carrier for an extended period. The sub-carriers are modulated in parallel contemporaneously, so that in combination the modulated carriers form an OFDM symbol. The OFDM symbol therefore comprises a plurality of sub-carriers each of which has been modulated contemporaneously with different modulation symbols. During transmission, a guard interval filled by a cyclic prefix of the OFDM symbol precedes each OFDM symbol. When present, the guard interval is dimensioned to absorb any echoes of the transmitted signal that may arise from multipath propagation. 
     It has been proposed for a television system known as the Advanced Television Systems Committee (ATSC) 3.0 in a publication entitled ATSC 3.0 Working Draft System Discovery and Signaling [1] to include a pre-amble in a transmitted television signal which is carrying broadcast digital television programmes. The preamble includes a so called “boot strap” signal which is intended to provide a receiver with a part of the transmitted signal which it can have a greater likelihood of detecting and therefore can serve as a signal for initial detection. This is because broadcasters anticipate providing multiple services, within a broadcast signal in addition to just broadcast television. Such services may be time-multiplexed together within a single RF channel. There is therefore a need to provide an easily detectable signal segment (the bootstrap signal) that is transmitted as part of a pre-amble to multiplexed frames, so that a receiver can discover and identify what signals and services are available. 
     It has been proposed [1] to make the bootstrap signal have a configuration, including sampling rate, signal bandwidth, subcarrier spacing, time-domain structure etc that is known to all receiver devices and to carry information to enable processing and decoding the wireless services associated with a detected bootstrap. This new capability ensures that broadcast spectrum can be adapted to carry new services and/or waveforms that are preceded by a universal entry point provided by the bootstrap for public interest to continue to be served in the future. 
     The bootstrap has been designed to be a very robust signal and detectable even at very low signal levels. As a result of this robust encoding, individual signalling bits within the bootstrap are comparatively expensive in terms of the physical resources that they occupy for transmission. Hence, the bootstrap is generally intended to carry only the minimum amount of information required for system discovery and for initial decoding of the following signal. 
     As can be appreciated, finding an efficient and cost effective technique for detecting the bootstrap signal represents a technical problem. 
     SUMMARY OF THE DISCLOSURE 
     Various further aspects and embodiments of the disclosure are provided in the appended claims, including a receiver for detecting and recovering payload data from a received signal. The received signal has been formed and transmitted by a transmitter to carry the payload data in Orthogonal Frequency Division Multiplexed (OFDM) symbols in one or more of a plurality of time divided frames, each frame including a preamble comprising a plurality of bootstrap OFDM symbols. One or more of the bootstrap OFDM symbols of the preamble carrying signalling data represented as a relative cyclic shift of a signature sequence which has been combined with the one or more of the bootstrap OFDM symbols. A bootstrap processor is configured to detect the signalling data from the bootstrap OFDM symbols using an estimate of the channel transfer function determined from one or more of the bootstrap OFDM symbols, and a demodulator circuit is configured to recover the payload data from the payload OFDM symbols using the signalling data. The bootstrap processor comprises a channel shaper, which is configured to convolve a time domain copy of the signature sequence with the estimate of the channel impulse response to generate a channel shaped signature sequence, a cross-correlator and a cyclic shift detector. A cross-correlator is configured to cross-correlate the useful part of each of the one or more bootstrap OFDM symbols with the regenerated and channel shaped copy of the signature sequence, and the cyclic shift detector is configured to estimate the signalling data conveyed by each of the one or more bootstrap OFDM symbols by detecting a cyclic shift of the signature sequence present in each of the one or more bootstrap OFDM symbols from a peak of the samples representing a result of the cross-correlation. 
     Embodiments of the present technique can decode the signalling data carried by the bootstrap OFDM symbols by identifying the cyclic shift of the signature sequence by cross-correlating the signature sequence with the bootstrap OFDM symbols in the time domain, which can provide a more efficient implementation. By cross-correlation in the time domain a more efficient decoding of the signalling data can be made, which requires only a single inverse Fourier transform. In some embodiments, the cyclic shift is detected in the frequency domain. 
     The present disclosure is supported by our co-pending patent applications numbers PCT/GB2014/050869, GB1305805.2, PCT/GB2014/050868, GB1305797.1, GB1305799.7, U.S. Ser. No. 14/226,937, PCT/GB2014/050870, GB1305795.5, PCT/GB2014/050954, GB1312048.0, TW103121570, PCT/GB2014/051679, EP13170706.9, PCT/EP2014/061467, GB1403392.2, GB1405037.1, TW103121568 and PCT/GB2014/051922, GB1420117.2 the entire contents of which are incorporated herein by reference. 
     Various further aspects and features of the present disclosure are defined in the appended claims, which include a method of transmitting payload data, a receiver and a method of detecting and recovering payload data. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Embodiments of the present disclosure will now be described by way of example only with reference to the accompanying drawings in which like parts are provided with corresponding reference numerals and in which 
         FIG. 1  provides a schematic diagram illustrating an arrangement of a broadcast transmission network; 
         FIG. 2  provides a schematic block diagram illustrating an example transmission chain for transmitting broadcast data via the transmission network of  FIG. 1 ; 
         FIG. 3  provides a schematic illustration of OFDM symbols in the time domain which include a guard interval; 
         FIG. 4  provides a schematic block of a typical receiver for receiving data broadcast by the broadcast transmission network of  FIG. 1  using OFDM; 
         FIG. 5  provides a schematic illustration of a sequence of transmission frames for transmitting broadcast data and payload data separated by a preamble carrying signalling data; 
         FIG. 6  provides a schematic representation of a preamble of one of the transmission frames shown in  FIG. 5 , which includes a so-called “bootstrap” signal or waveform comprised of multiple OFDM symbols; 
         FIG. 7  provides a schematic block diagram of a part of the transmitter shown in  FIG. 2  for transmitting a bootstrap signal comprising a plurality of bootstrap, OFDM symbols; 
         FIG. 8  provides a schematic representation of a bootstrap OFDM symbol in the frequency domain; 
         FIG. 9  is an illustrative flow diagram representing the operation of the transmitter in imprinting the signalling data, which is transported on one or more of the bootstrap OFDM symbols by cyclically shifting the time domain symbol sequence; 
         FIG. 10  provides a schematic representation of a time domain structure of a first of the bootstrap OFDM symbols; 
         FIG. 11  provides a schematic representation of a second time domain structure of one or more other bootstrap OFDM symbols; 
         FIG. 12  is a schematic block diagram of an example receiver for detecting and recovering signalling from the one or more bootstrap OFDM symbols; 
         FIG. 13  is a schematic block diagram of a receiver for detecting a bootstrap OFDM symbol including identifying a trigger time for performing a forward Fourier transform on an OFDM symbol; 
         FIG. 14  is a more detailed example of the CAB detector of  FIG. 14 , which includes a detector for identifying the triggering time for performing a forward Fourier transform from a first of the bootstrap OFDM symbols having the first time domain structure of  FIG. 10 ; 
         FIG. 15  is a schematic block diagram of a bootstrap OFDM symbol processor for estimating the cyclic shift that was used by the transmitter to imprint the signalling data on to one or more detected bootstrap OFDM symbols; 
         FIG. 16  is a schematic block diagram of an example receiver for detecting bootstrap OFDM symbols which includes a combined bootstrap signal detector for identifying a trigger time for performing a forward Fourier transform on an OFDM symbol according to the present technique; 
         FIG. 17  is a schematic block diagram of a bootstrap signal detector, which is configured to provide an improved estimate of an identified trigger time for performing an FFT on received OFDM symbols from four OFDM symbols of a bootstrap signal in accordance with the present technique; 
         FIG. 18  is a schematic block diagram providing an example embodiment of a first bootstrap processor/decoder in accordance with the present technique; 
         FIG. 19  is a schematic block diagram of a polyphase processor forming part of the first bootstrap processor/decoder showing in  FIG. 18 ; 
         FIG. 20  is a schematic representation of a process of decimating a received signal and multiplying the received signal by the coefficients of a signature sequence in accordance with the present technique; 
         FIG. 21  represents a process of correlating a received OFDM symbol in the frequency domain with a reproduced signature sequence performed by the first bootstrap processor/decoder; 
         FIG. 22  is a schematic block diagram of a second bootstrap processor/decoder for detecting signalling data carried by other bootstrap symbols other than the first bootstrap OFDM symbol; 
         FIG. 23  is a schematic block diagram of parts of the second bootstrap processor/decoder illustrated in  FIG. 22  which is configured to estimate the cyclic shift of a signature sequence carried by the bootstrap symbol to detect signalling data in accordance with the present technique; 
         FIG. 24  is a schematic representation of a process of correlating the signature sequence with a received signal using decimation in accordance with the present technique; 
         FIG. 25  is a schematic block diagram of an alternative embodiment for the second bootstrap processor/decoder illustrated in  FIGS. 22 and 23  in which the signalling data is detected from an estimate of the cyclic shift applied to the signature sequence; 
         FIG. 26  is an illustrative flow diagram representing an operation of a first bootstrap processor in accordance with the present technique; and 
         FIG. 27  is a representative flow diagram illustrating a method of operation of a second bootstrap processor in accordance with the present technique. 
     
    
    
     DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS 
     Embodiments of the present disclosure can be arranged to form a transmission network for transmitting signals representing data including video data and audio data so that the transmission network can, for example, form a broadcast network for transmitting television signals to television receiving devices. In some examples the devices for receiving the audio/video of the television signals may be mobile devices in which the television signals are received while on the move. In other examples the audio/video data may be received by conventional television receivers which may be stationary and may be connected to a fixed antenna or antennas. 
     Television receivers may or may not include an integrated display for television images and may be recorder devices including multiple tuners and demodulators. The antenna(s) may be inbuilt to television receiver devices. The connected or inbuilt antenna(s) may be used to facilitate reception of different signals as well as television signals. Embodiments of the present disclosure are therefore configured to facilitate the reception of audio/video data representing television programs to different types of devices in different environments. 
     As will be appreciated, receiving television signals with a mobile device while on the move may be more difficult because radio reception conditions will be considerably different to those of a conventional television receiver whose input comes from a fixed antenna. 
     An example illustration of a television broadcast system is shown in  FIG. 1 . In  FIG. 1  broadcast television base stations  1  are shown to be connected to a broadcast transmitter  2 . The broadcast transmitter  2  transmits signals from base stations  1  within the coverage area of the broadcast network. The television broadcast network shown in  FIG. 1  may operate as a so called multi-frequency network where each television broadcast base station  1  transmits its signal on a different frequency than other neighbouring television broadcast base stations  1 . The television broadcast network shown in  FIG. 1  may also operate as a so called single frequency network in which each of the television broadcast base stations  1  transmit the radio signals conveying audio/video data contemporaneously so that these can be received by television receivers  4  as well as mobile devices  6  within the coverage area of the broadcast network. For the example shown in  FIG. 1  the signals transmitted by the broadcast base stations  1  are transmitted using Orthogonal Frequency Division Multiplexing (OFDM) which can provide an arrangement for transmitting the same signals from each of the broadcast stations  2  which can be combined by a television receiver even if these signals are transmitted from different base stations  1 . Provided a spacing of the broadcast base stations  1  is such that the propagation time between the signals transmitted by different broadcast base stations  1  is less than or does not substantially exceed a guard interval that precedes the transmission of each of the OFDM symbols then a receiver device  4 ,  6  can receive the OFDM symbols and recover data from the OFDM symbols in a way which combines the signals transmitted from the different broadcast base stations  1 . Examples of standards for broadcast networks that employ OFDM in this way include DVB-T, DVB-T2 and ISDB-T. 
     An example block diagram of a transmitter forming part of the television broadcast base stations  1  for transmitting data from audio/video sources is shown in  FIG. 2 . In  FIG. 2  audio/video sources  20  generate the audio/video data representing television programmes. The audio/video data is encoded using forward error correction encoding by an encoding/interleaver block  22  which generates forward error correction encoded data which is then fed to a modulation unit  24  which maps the encoded data onto modulation symbols which are used to modulate OFDM symbols. Depicted on a separate lower arm, signalling data providing physical layer signalling for indicating for example the format of coding and modulation of the audio/video data is generated by a physical layer signalling unit  30  and after being encoded by an encoding unit  32 , the physical layer signalling data is then modulated by a modulation unit  24  as with the audio/video data. 
     A frame builder  26  is arranged to form the data to be transmitted with the physical layer signalling data into a frame for transmission. The frame includes a time divided section having a preamble in which the physical layer signalling is transmitted and one or more data transmission sections which transmit the audio/video data generated by the audio/video sources  20 . An interleaver  34  may interleave the data which is formed into symbols for transmission by an OFDM symbol builder  36  and an OFDM modulator  38 . The OFDM symbol builder  36  receives pilot signals which are generated by a pilot and embedded data generator  40  and fed to the OFDM symbol builder  36  for transmission. The output of the OFDM modulator  38  is passed to a guard insertion unit  42  which inserts a guard interval and the resulting signal is fed to a digital to analogue convertor  44  and then to an RF front end  46  before being transmitted by an antenna  48 . 
     As with a conventional arrangement OFDM is arranged to generate symbols in the frequency domain in which data symbols to be transmitted are mapped onto sub carriers which are then converted into the time domain using an inverse Fourier Transform which may comprise part of the OFDM modulator  38 . Thus the data to be transmitted is formed in the frequency domain and transmitted in the time domain. As shown in  FIG. 3  each time domain symbol is generated with a useful part of duration Tu seconds and a guard interval of duration Tg seconds. The guard interval is generated by copying a part of the useful part of the symbol with duration Tg in the time domain, where the copied part may be from an end portion of the symbol. By correlating the useful part of the time domain symbol with the guard interval, a receiver can be arranged to detect the start of the useful part of the OFDM symbol which can be used to trigger a Fast Fourier Transform to convert the time domain symbol samples into the frequency domain from which the transmitted data can then be recovered. Such a receiver is shown in  FIG. 4 . 
     In  FIG. 4  a receiver antenna  50  is arranged to detect an RF signal which is passed via a tuner  52  and converted into a digital signal using an analogue to digital converter  54  before the guard interval is removed by a guard interval removal unit  56 . After detecting the optimum position for performing a fast Fourier Transform (FFT) to convert the time domain samples into the frequency domain, an FFT unit  58  transforms the time domain samples to form the frequency domain samples which are fed to a channel estimation and correction unit  60 . The channel estimation and correction unit  60  estimates the transmission channel used for equalisation for example by using pilot sub-carriers which have been embedded into the OFDM symbols. After excluding the pilot sub-carriers, all the data-bearing sub-carriers are fed to a de-mapper unit  62  which extracts the data bits from the sub-carriers of the OFDM symbol. These data bits are then fed to a de-interleaver  64  which de-interleaves the sub-carrier symbols. The data bits are now fed to a bit de-interleaver  66 , which performs the de-interleaving so that the error correction decoder can correct errors in accordance with a conventional operation. 
     Framing Structure 
       FIG. 5  shows a schematic diagram of the framing structure of a frame that may be transmitted and received in the systems described with reference to  FIGS. 1 to 4 .  FIG. 5  illustrates different physical layer frames, some targeted for mobile reception whilst others are targeted for fixed roof-top antenna reception. The system can be expanded in future to incorporate new types of frames, for the current system, these potential new types of frames are simply known as future extension frames (FEFs). 
     The framing structure shown in  FIG. 5  is therefore characterised by frames which may each include payload data modulated and encoded using different parameters. This may include for example using different OFDM symbol types having different number of sub-carriers per symbol, which may be modulated using different modulation schemes, because different frames may be provided for different types of receivers. However each frame may include at least one OFDM symbol carrying signalling data, which may have been modulated differently to the one or more OFDM symbols carrying the payload data. Furthermore for each frame, the signalling OFDM symbol may be a different type to the OFDM symbol(s) carrying the payload data. The signalling data is required to be recovered so that the payload data may be de-modulated and decoded. 
     Bootstrap Signal 
     As explained in [1], the bootstrap signal provides a universal entry point into an ATSC way form. The bootstrap signal is supposed to have a known configuration in that the sampling rate, the signal bandwidth, the sub carrier spacing and time domain structure are known a priori at the receivers.  FIG. 6  provides a schematic representation of the form of a bootstrap signal with respect to the data carrying frames shown in  FIG. 5 . In  FIG. 6  the bootstrap signal, which may form part of the preamble  104 ,  106 ,  108 ,  110 , precedes a data-bearing frame  100 ,  102 ,  112 . The bootstrap signal comprises four or more OFDM symbols beginning with a synchronisation symbol positioned at the start of each frame to enable service discovery, coarse time synchronisation, frequency offset estimation and initial channel estimation. The remaining other bootstrap OFDM symbols contain sufficient control signalling to provide communications parameters to allow the received signal to be decoded for the remaining part of the frame. Thus the bootstrap signal carries signalling information to enable a receiver to discover the parameters with which the data-bearing frame has been configured so that a receiver can detect and recover this data. 
     A schematic block diagram of a part of the transmitter shown in  FIG. 2  which is configured to transmit a bootstrap signal is shown in  FIG. 7 . In  FIG. 7  a signature sequence generator  700  is arranged to generate a signature sequence which is mapped onto the sub carriers of an OFDM symbol forming the bootstrap symbol by the sub carrier mapping and zero padding unit  702 . The frequency domain signal is then transformed into the time domain by an inverse Fourier transform  704 . Signalling information which is to be transmitted with the bootstrap signal is fed on a first input  705  to a cyclic shift unit  706 . The cyclic shift unit  706  also receives on a second input  707  the time domain OFDM representing the bootstrap symbol. As will be explained below, signalling information is represented as a cyclic shift of the bootstrap OFDM symbol in the time domain. The cyclically shifted bootstrap OFDM symbol is then fed to a guard interval insertion unit  708 , which adds a guard interval to the bootstrap OFDM symbol in the form in which the OFDM symbol forming of the bootstrap symbol will be transmitted by a transmitter unit  709 . 
     As shown in  FIG. 7  the signature sequence generator  700  generates a signature sequence comprising a pseudo random sequence generator  710  and a Zadoff-Chu sequence generator  712 . These two sequences are multiplied together by a multiplier  714  before the combined sequences are mapped onto the sub carriers of the OFDM symbol by the sub carrier mapping and zero padding unit  702 . As shown in  FIG. 7  the seed value for the pseudo random number generator  710  is fed on a first input  720  and a second input  722  provides an indication of the route of the Zadoff-Chu sequence generator  712 . 
     The mapping of the Zadoff-Chu (ZC) sequence modulated by a pseudo random sequence or pseudo noise (PN) to form the signature sequence onto the OFDM symbol in a symmetrical way is shown in  FIG. 8 . 
     As shown in  FIG. 8 , in the frequency domain, the bootstrap signal can be regarded as comprising two halves  810  of a symmetrical Zadoff-Chu (ZC) sequence. Each symbol in the Zadoff-Chu sequence is arranged to modulate an active carrier  812 . Correspondingly the PN sequence is arranged to modulate the sub-carriers as shown by the lines  814 . Other sub carriers of the bootstrap symbol are not used and so are set to zero as shown for example at either end of the bootstrap signal  820 ,  822 . 
     As shown in  FIG. 8  the ZC sequence and the PN sequence are mapped to the OFDM sub-carriers in a manner that produces a reflective symmetry about the central DC sub carrier of the OFDM symbol. The subcarrier values for the n-th symbol of the bootstrap (0≦n&lt;N B ) may be calculated as in the following equation, where N H =(N ZC −1)/2, N B  is the number of bootstrap symbols and p(k) are elements of the PN-sequence The ZC sequence determined by its root q, may be the same for each symbol, while the PN sequence shall advance with each symbol. 
     
       
         
           
             
               
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     In one example, signalling data can be conveyed in the bootstrap signal by performing a data-determined cyclic shift of the symbol in the time domain. This is performed by the cyclic shift blocks shown in  FIG. 7 . The process for conveying signalling bits is summarised in  FIG. 9 . 
     In  FIG. 9  the frequency domain sequence is formed by the sequence generator  700  in the frequency domain in step S 900 . In step S 902  an inverse Fourier transform is performed by the IFFT module  704  to convert the frequency domain signal into the time domain. Thus in step S 904  the sequence is formed in the time domain. As shown in step S 906  the signalling bits are formed and then interpreted in step S 908  as a relative cyclic shift value and in step S 910  the relative shift value is converted to an absolute shift value. As shown by arrow S 912  the time domain sequence formed in step S 904  is then shifted in accordance with the absolute cyclic shift determined in step S 910 . Finally, in step S 914  the time domain sequence to be transmitted is produced. 
     Time Domain Structure 
     Each of the bootstrap OFDM symbols can be interpreted as being comprised of three parts which are referred to as A, B and C. As explained above, an OFDM symbol is usually formed with a guard interval generated by copying a section of the OFDM symbol in the time domain as a preamble to the OFDM symbol in order to account for multi path reception at the receiver. Each bootstrap symbol is formed in one of two ways. The different formation of the bootstrap symbols in the time domain is shown in  FIGS. 10 and 11 . As shown in both  FIGS. 10 and 11  the data carrying part of the symbol that is the original formation of the OFDM symbol before guard intervals are added is represented as section A. Thus, section A is derived as the IFFT of the frequency domain structure with or without the cyclic shift explained above to represent the signalling bits being conveyed by the bootstrap symbol. Parts B and C are composed of samples taken from the end of A with a frequency shift of ±f Δ  which is equal to the sub carrier spacing introduced into the samples of B by the transmitter, and correspondingly removed at the receiver. Each bootstrap symbol consistently consists of 3072 samples. 
     There are two variations of the time domain structure of the bootstrap symbols which are referred to as C−A−B and B−C−A. The initial symbol of the bootstrap referred to as bootstrap symbol zero is provided for synchronisation detection and employs the C−A−B structure which is shown in  FIG. 10  and applies a frequency shift of +f Δ  to part B. The remaining bootstrap symbols use the B−C−A structure including the final bootstrap symbol with a phase inversion which provides the termination of the bootstrap signal as explained above and applies a frequency shift of −f Δ  to part B. 
     Bootstrap Detector 
     A schematic block diagram illustrating an adaptation of the receiver shown in  FIG. 4  when operating to detect the bootstrap signal is shown in  FIG. 12 . As shown in  FIG. 4  the signal detected by an antenna  50  is fed to an RF tuner  52  and then to an A to D converter  54 . The received digitally sampled signal is then fed to a Forward Fourier Transform processor  58  and also to a first input of a switch  1201 , which is controlled by a controller  1202  to switch the received digitally sampled signal between a bootstrap detector  1204  and a second of two bootstrap processors  1206 ,  1210 . The bootstrap detector  1204  generates a trigger signal fed on a channel  1208  to the FFT processor  58  in order to identify a most useful part of the received signal which is to be converted from the time to the frequency domain to validate the bootstrap signal and to recover the signalling data. An output of the FFT processor  58  provides a frequency domain version of the received signal to a first of the bootstrap processors  1210 . The first bootstrap processor  1210  is configured to generate a first estimate of the channel transfer function (CTF) H(z) at an output channel  1212 . 
     Example detectors of the bootstrap symbols of the bootstrap signal are provided in  FIGS. 13, 14 and 15 . As indicated above only the first bootstrap symbol has the C−A−B structure which is transmitted to provide initial synchronisation.  FIG. 13  provides an example block diagram of a detector for the first bootstrap symbol. As shown in  FIG. 13  the received discrete time signal r (n) is fed to a delay unit  1301  and a C−A−B structure detector  1302 . The C−A−B structure detector  1302  will be explained in more detail with reference to  FIG. 14  shortly. The C−A−B structure detector  1302  generates on a first output  1304  an estimate of a fine frequency offset (FFO), which is a frequency shift smaller than the OFDM symbol sub-carrier spacing and which may have occurred during the transmission of the bootstrap OFDM symbol. Also output from a second channel  1306  is an indication of a timing trigger for indicating the period of the received OFDM symbol which is transformed by the FFT processor  58 , so as to capture as far as possible a maximum amount of energy of the received OFDM bootstrap symbol. However before transforming the received bootstrap symbol into the frequency domain, a total frequency offset is removed by a multiplier  1308 . The multiplier  1308  receives on a first input the delayed received signal from the delay unit  1301  and on a second input an inverse of a total frequency offset which is formed by an adder  1310  and a tone generator  1312 . The total frequency offset is formed by the adder  1310  from at least one of the fine frequency offset (FFO) estimated by the C−A−B detector  1302  which is fed to a first input and an integer frequency offset (IFO) estimated by the bootstrap signal processor  1310 . This total frequency offset is input into the tone generator  1310  causing it to generate a sinusoidal tone at a frequency equal to the total frequency offset. The bootstrap signal processor  1210  generates the IFO by correlating the frequency domain sub-carriers with a re-generated version of the signature sequence generated from a combination of the ZC sequence modulated with the PN sequence. The location of a peak of the correlation output is then used to estimate the IFO, which is a displacement in the frequency domain of a number of sub-carriers with respect to a frequency reference within the frequency band of the bootstrap signal. Thus the total frequency offset is estimated and removed by the multiplier  1308  and the tone generator  1210  from the FFO estimated by the CAB structure detector  1302  and the IFO estimated by the bootstrap signal processor  1210 . 
     As indicated above the detector  1302  shown in  FIG. 13  for detecting the first bootstrap OFDM symbol is used to generate a FFO and indicate the useful part of the input signal burst for Forward Fourier Transform (FFT). The detector  1302  shown in  FIG. 13  is shown in more detail in  FIG. 14 . As shown in  FIG. 14 , one example of the C−A−B detector  1302  is provided with three moving averaging filters  1401 ,  1402 ,  1403  which are fed from the output of multipliers  1410 ,  1412 ,  1414 . The received signal is input on a channel  1420  and fed respectively to three delay circuits  1422 ,  1424 ,  1426  which serve to delay the received signal by a number of samples respectively equal to the number in the A, A+B and B parts of the bootstrap symbol. A multiplier  1428  frequency shifts the received signal which is input to the second and third delay portions  1424 ,  1426  by a frequency shift corresponding to the frequency adjustment ∓Δf fed from a tone generation circuit  1430 . After respectively delaying the received signals by the number of samples in the A, A+B and B parts of the bootstrap symbol, the outputs respectively from the delay elements  1422 ,  1424 ,  1426  are fed to the multipliers  1410 ,  1412 ,  1414 . The received signal is then multiplied by the multipliers  1410 ,  1412 ,  1414  from the output from the delay elements  1422 ,  1424 ,  1426  with a conjugated sample of the received signal samples to form in combination with the moving averaging filters  1401 ,  1402 ,  1403  a correlation of the received signal with respect to itself following the delay A, A+B and B. The outputs of the moving average filters  1401 ,  1402 ,  1403  are delayed by delay elements  1440  and up-scaled by scaling elements  1442 ,  1444  and summed by an adder  1450  to generate a peak combined sample by correlating each of the sections C, A and B of the received signal with their respective copies to identify a peak at which the FFT trigger point is detected at the output  1460 . Correspondingly, the phase of the peak determines the FFO provided on the output  1462 . 
     As shown also in  FIG. 14  the moving average filter is formed from a delay element  1480  whose output is subtracted from its input and then added to a previous output to form a moving average sample correlation value. 
       FIG. 15  provides a schematic block diagram of a bootstrap decoder for decoding the second, third and fourth and any other following bootstrap OFDM symbols. As shown in  FIG. 15  the received signal r(n) is received by a delay unit  1301  and fed to a multiplier  1308  which multiplies the signal received from the delay unit  1301  by a tone signal provided by a tone generation unit  1312 . The total frequency offset is formed from adding the IFO fed to a first input  1502  to an adder  1310  with a second input  1304  being provided with an indication of the FFO. Both the FFO and the IFO may be generated by the bootstrap processor of the first bootstraps OFDM symbol shown in  FIGS. 13 and 14  as explained above. The received signal which is delayed by N S =3072 samples and after the total frequency offset has been removed is then fed to a B−C removal unit  1520 . The B−C removal unit  1520  isolates the useful portion A of the bootstrap symbol which is then fed to a first input  1522  of a relative cyclic shift estimation unit  1524 . 
     A first input  1532  to a multiplier  1530  receives an estimate of the channel H(z) in the frequency domain and a second input  1536  receives a signature sequence provided from a look-up table  1534 . The multiplier forms at an output  1538  a multiplication of the channel frequency response and the signature sequence in the frequency domain, which is equivalent to convolution with the channel impulse response in the time domain. The output  1538  is therefore a channel-shaped version of the signature sequence (ZC sequence combined with PN sequence), which is fed to an inverse Fourier transform processor  1540  which converts the channel-shaped signature sequence into the time domain which is fed to a second input  1542  of the relative cyclic shift estimation unit  1524 . 
     The relative cyclic shift estimation unit  1524  then cyclically correlates the received useful portion of the first bootstrap OFDM symbol with the channel-shaped signature sequence to identify a relative cyclic shift of the signature sequence as formed at the transmitter. The output  1550  of the relative cyclic shift estimation unit  1524  is fed to an adder  1552  which subtracts from the cyclic shift detected from the previous symbol. Accordingly an absolute cyclic shift decoding unit  1560  is used to decode the total cyclic shift on the received OFDM symbol which is then output as M n  on an output channel  1562 . 
     The example shown in  FIG. 15  provides an example of a time domain correlator which is configured to detect a cyclic shift of the signature sequence and therefore the signalling information carried by the bootstrap symbol n, where n&gt;1. A more detailed explanation of the example shown in  FIG. 15  will be given below. 
     Combined Correlator 
     Embodiments of the present technique can provide an arrangement in which the separate detectors for the time delayed structure of the OFDM bootstrap symbols are combined in order to increase an accuracy with which the FFT trigger point and fine frequency offset is determined. As explained above, a conventional arrangement only uses the first OFDM bootstrap symbol to detect the FFO and the FFT trigger point. However embodiments of the present technique are arranged to combine all of the bootstrap symbols to form an estimate of the FFT trigger point and the fine frequency offset with improved accuracy. An example embodiment is shown in  FIG. 16 . 
       FIG. 16  corresponds to the bootstrap receiver shown in  FIG. 12  but has been adapted in accordance with the present technique to provide improvements in detecting the bootstrap signal. Therefore, only differences between  FIGS. 12 and 16  will now be described. 
     As shown in  FIG. 16  the received signal r(n) is fed from the output of the A/D convertor  54  to an input of a combined bootstrap detector  1601 . As we explain in more detail below with reference to  FIG. 17 , the combined bootstrap detector  1601  uses all of the bootstrap OFDM symbols to detect the FFT trigger point fed on the output channel  1208 . Furthermore, in contrast to the bootstrap receiver shown in  FIG. 12 , the arrangement shown in  FIG. 16  shows a bootstrap processor for the first OFDM symbol  1602  configured to generate an indication of the IFO on a first channel  1604  and an indication of the FFO on a second channel  1606 . Both channels  1604 ,  1606  providing an indication of the IFO and the FFO respectively are fed to the second bootstrap processor  1610  which decodes the second, third and fourth bootstrap OFDM symbols to generate an indication of the signalling information M n  communicated by these OFDM symbols. 
     As shown in  FIG. 17 , a bootstrap signal detector according to an embodiment of the present technique is arranged to detect and to recover a timing for the FFT trigger and the FFO from the two types of time domain structures of the OFDM symbols of the bootstrap signal. Accordingly, a mirrored structure about the received input channel  1701  is provided for correlating the B−C−A correlator  1705  and the C−A−B correlator  1703  respectively. Each of the respective correlators forms a correlation sum for the respective structures of the OFDM bootstrap symbols. The first correlator  1703  is the C−A−B correlator that is used for detecting that first bootstrap symbol. This has already been described above. The difference here is that the output of the last adder  1742  is delayed by a number of samples equivalent to the duration of all the other bootstrap symbols i.e. a total of three times A+B+C samples time in a system with four bootstrap symbols before being fed to an input of a final adding (combiner) stage  1744  which combines the results from all the B−C−A correlators pertaining to the rest of the bootstrap symbols 
     Correspondingly in respect of the lower correlator for the B−C−A branch, delay elements  1750 ,  1752 ,  1754  form in combination with the multipliers  1756 ,  1758 ,  1760  a multiplication of each of the samples of the received signal conjugated and multiplied by a corresponding delayed sample delayed respectively by the number of samples in the A, A+B, A part of the bootstrap symbol. As with the first branch a tone generator  1762  offsets the frequency of the lower two branches which are multiplied by multipliers  1764  to adjust the frequency of the lower two branches by −f Δ . The moving average filters  1766 ,  1768 ,  1770  serve to perform a moving average of the multiplier output samples which are then fed via gain units  1772 ,  1774  to an adder  1776 . The outputs are then fed to the adder  1744  representing the correlation contribution from the last bootstrap symbol. The output of adder  1776  is also delayed by an amount which is equivalent to the total number of samples in one bootstrap OFDM symbol. The output of delay element  1780 , representing the correlation contribution from the third bootstrap symbol, is also fed to the adder  1744 . The output of delay element  1780  is in turn also delayed by an amount which is equivalent to the total number of samples in one bootstrap OFDM symbol using the delay element  1782  whose output is then also fed as the correlation contribution from the second bootstrap symbol, to the adder  1744 . Thus in combination the two branches  1703 ,  1705  serve to combine the correlation sums for all the OFDM symbols on the bootstrap signal to generate as an output  1790  a combined estimate of the optimum trigger point and an estimate of the fine frequency offset as an output  1792 . 
     As for the first correlator  1703 , for the lower correlator  1705 , the first delay element  1750 , the first multiplier  1756  and the first moving averaging filter  1766  together form a first auto-correlator for a part A of the one or more other bootstrap OFDM symbols. The second delay element  1752 , the second multiplier  1756  and the second moving averaging filter  1768  together form a second auto-correlator for a part C of the bootstrap OFDM symbol and the third delay element  1754 , the third multiplier  1760  and the third moving averaging filter  1770  together form a third auto-correlator for a part B of the bootstrap OFDM symbol. 
     As will be appreciated, although in  FIG. 16  the first bootstrap processor  1602  for processing the first bootstrap symbol and the second bootstrap processor  1610  for processing the one or more other bootstrap symbols are shown as separate processors, in other examples the first and second processors  1602 ,  1610  may be configured as the same bootstrap processor. 
     Bootstrap Processor for First OFDM Symbol 
     A more detailed explanation of the features and operation of the first bootstrap processor shown in  FIG. 16  will now be explained with reference to an example block diagram shown in  FIG. 18 . As shown in  FIG. 18  the frequency domain samples representing the sub-carriers of the received OFDM symbol are fed to an up-sampling unit  1802 . The frequency domain OFDM symbol is then up sampled in accordance with a factor U. The factor U maybe anyone or for example 4, 8, 16 so that if the OFDM symbol has 2048 sub-carriers then the number samples is increased to 8 k, 16 k or 32 k. In some embodiments, the up-sampling is achieved by writing the 2048 samples in the FFT window to the initial samples of an 8 k, 16 k or 32 k FFT input buffer for which the rest of the samples are set to zero. When the 8 k, 16 k or 32 k FFT is executed, its output is respectively a 4, 8 or 16 times up-sampling of the symbol spectrum. The up-sampled frequency domain OFDM symbol is then fed to a cross-correlator  1804 . The cross correlator  1804  receives on a second input  1806  samples corresponding to one of the signature sequences which have been pre-stored in the data store  1808 . Each pre-stored sequence represents the Zadoff-Chu sequence multiplied by one version of the PN sequence, which is fed to the cross correlator  1804 . Each version of the PN sequence is generated using the same polynomial but with a different starting seed. 
     In one example the up-sampling unit  1802  is configured to form the up-sampled frequency domain version of the bootstrap OFDM symbol, by identifying samples of the bootstrap symbol in the time domain using the synchronisation timing, appending a plurality of zero valued samples to the samples of the bootstrap OFDM symbol, and performing a forward Fourier transform to convert the bootstrap OFDM symbol with the appended zero valued samples into the frequency domain, the number of zero valued samples being (U−1) times the number of samples of the bootstrap OFDM symbol in the time domain. 
     In accordance with the present technique the cross correlator  1804  correlates the up-sampled version of the OFDM symbol in turn with each of the base band sampled signature sequences to generate at an output  1812  samples of the cross correlation between the signature sequence and the up-sampled OFDM symbol. The signature sequence which provides the maximum cross-correlation is deemed to be the one used at the transmitter. The cross-correlation output from this signature sequence is then processed further. 
     A peak detector  1814  then detects the sample position of the cross correlation peak and so provides this indication to an output processor  1816 . The output processor  1816  processes the position of the peak in the cross correlation output to identify, the signature sequence, the IFO and FFO which are output on channels  1820 ,  1822 . 
     The cross-correlator therefore calculates the cross-correlation sum: 
     
       
         
           
             
               F 
               ⁡ 
               
                 ( 
                 l 
                 ) 
               
             
             = 
             
                
               
                 
                   ∑ 
                   
                     k 
                     = 
                     0 
                   
                   
                     k 
                     = 
                     
                       
                         N 
                         ZC 
                       
                       - 
                       1 
                     
                   
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     
                       C 
                       1 
                       * 
                     
                     ⁡ 
                     
                       ( 
                       k 
                       ) 
                     
                   
                   ⁢ 
                   
                     R 
                     ⁡ 
                     
                       ( 
                       
                         k 
                         + 
                         l 
                       
                       ) 
                     
                   
                 
               
                
             
           
         
       
     
     where l=−E, −E+1, . . . 0, 1, E−1, E is the number of edge subcarriers in the boostrap symbols, so that for a 2 k OFDM symbol, then E=ceil((N FFT −N ZC )/2)=(ceil(2048−1499)/2)=275. 
     The above calculation requires 2N ZC E complex multiplies and 2N ZC E complex adds. 
     A potential simplification for the cross-correlator would be to take advantage of the symmetry of the signature sequence, and so perform the cross-correlation according to the following equation. 
     
       
         
           
             
               F 
               ⁡ 
               
                 ( 
                 l 
                 ) 
               
             
             = 
             
                
               
                 
                   ∑ 
                   
                     k 
                     = 
                     0 
                   
                   
                     k 
                     = 
                     
                       
                         ( 
                         
                           
                             N 
                             ZC 
                           
                           - 
                           1 
                         
                         ) 
                       
                       / 
                       2 
                     
                   
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     
                       C 
                       1 
                       * 
                     
                     ⁡ 
                     
                       ( 
                       k 
                       ) 
                     
                   
                   ⁢ 
                   
                     ( 
                     
                       
                         R 
                         ⁡ 
                         
                           ( 
                           
                             k 
                             + 
                             l 
                           
                           ) 
                         
                       
                       + 
                       
                         R 
                         ⁡ 
                         
                           ( 
                           
                             
                               N 
                               ZC 
                             
                             - 
                             1 
                             - 
                             k 
                             - 
                             l 
                           
                           ) 
                         
                       
                     
                     ) 
                   
                 
               
                
             
           
         
       
     
     This calculation requires only N ZC E complex multiplies and 2N ZC E complex adds, and therefore has an advantage of the reducing the complexity of calculating the cross-correlation. 
     The signature sequence identified may be one out of eight possible sequences. From the FFT sample or frequency bin identified by the output processor  1816 , the IFO can be identified from l[max (F(l))]×Δf—where l is FFT bin which shows the peak cross-correlation and the Δf is the sub-carrier bandwidth of the bootstrap. In some embodiments, Δf=3000 Hz. 
     As explained above, according to an example embodiment of the present technique, the up-sampler  1802  is configured to up-sample the frequency domain version of the OFDM symbol by a factor U. Accordingly the cross-correlator  1804  performs a cross correlation between the received OFDM symbol and the signature sequence in the frequency domain with the received OFDM symbol being up-sampled by a factor U so that the cross-correlation becomes as follows: 
     
       
         
           
             
               F 
               ⁡ 
               
                 ( 
                 
                   Ul 
                   + 
                   i 
                 
                 ) 
               
             
             = 
             
                
               
                 
                   ∑ 
                   
                     k 
                     = 
                     0 
                   
                   
                     k 
                     = 
                     
                       
                         N 
                         ZC 
                       
                       - 
                       1 
                     
                   
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     
                       C 
                       1 
                       * 
                     
                     ⁡ 
                     
                       ( 
                       k 
                       ) 
                     
                   
                   ⁢ 
                   
                     R 
                     ⁡ 
                     
                       ( 
                       
                         
                           U 
                           ⁡ 
                           
                             ( 
                             
                               l 
                               + 
                               k 
                             
                             ) 
                           
                         
                         + 
                         i 
                       
                       ) 
                     
                   
                 
               
                
             
           
         
       
     
     Then according to the present technique, the output processor  1816  is configured to detect the IFO and the FFO according to the following expressions: 
     Let l′=l[max (|F(l)|)] then
 
IFO=round( l′÷U )×Δ f  
 
FFO=(Δ f l′÷U )−IFO
 
     The identified signature sequence for the first OFDM bootstrap symbol C 1 (k) is then fed to a divider  1830  which receives on a second input  1832  the frequency domain OFDM symbol. By dividing the received signal by the signature sequence for the first bootstrap OFDM symbol an estimate of the channel transfer function H(k) for the first OFDM bootstrap symbol is generated at an output  1834 . 
     A more detailed block diagram of the cross correlator  1804  is shown in  FIG. 19 . As shown in  FIG. 19  the signature sequence C 1 (k) is fed on a first input  1806  to a conjugator  1901 . The up-sampled OFDM symbol is fed on an input  1902  to a sequence of down sampling units  1904  which successively extract one sample from every U of the up sampled version of the OFDM symbol which are fed to a first input of a multiplier  1906 . The received signal is also fed via a sequence of delay units  1908  to successive down samplers  1904  so that successively each of the samples of the up sampled version of the OFDM symbol are multiplied respectively by different samples of the signature sequence in the base band domain which is fed from the conjugator  1901  on channels  1910 . Each of the multipliers feeds its output to a moving averaging filter  1912  of length N ZC . Each of the outputs of a moving average filters  1912  are then fed to a terminal  1914  which forms part of the demultiplexer  1916  which selects in turn the output from each of the moving average filters  1912  in order to output the result of the cross-correlator  1804  which forms the input of the peak detector  1814 . 
     The cross correlator  1804 , which performs the correlation of the up-sampled version of the OFDM symbol with the signature sequence in order to identify the IFO and the FFO from a peak of the correlation is represented pictorially in  FIG. 20 . A first section  2001  represents the samples of the signature sequence C(k) in this example only six samples are shown (for k=0 to 5). Correspondingly, the up-sampled version of the received OFDM symbol is shown in section  2002 . In this example, the up-sampling facture U is set to 4. As shown in the first row  2004 , each of the samples of the signature sequence  2001  are multiplied by one of the samples of the up-sampled version of the received OFDM symbol  2002  for a first literation l=0. Further rows  2006 ,  2008 ,  2010  are shown which correspondingly show a next of the up-sampled version of the OFDM symbol being multiplied by the corresponding co-efficient of the signature sequence. Accordingly, in turn, sets of the samples representing the sub-carriers of the OFDM symbol R(Ul) for the over sampled version are successively multiplied by the coefficients C(k) of the signature sequence, to identify one of those sets of samples which produces the peak in the correlation result. According to this poly-phase processing of the cross-correlation, the IFO can be identified from a peak of the sub-carrier which produces the highest correlation result and the peak position with respect to the over-sampled sub-carriers identifies the FFO. According to the present technique therefore a more accurate estimate of both the FFO and the IFO is provided. 
     A further example function of the cross correlator shown in  FIGS. 18 and 19  is represented in  FIG. 21 . In  FIG. 21 , a correlation of the signature sequence  2101  with the OFDM symbol  2102  over a range represented by two extremes shown in  FIG. 21  is shown for a range which exceeds the edge sub-carriers E as explained above. Accordingly, by extending the range of cross correlation the range of IFO that can be estimated also increases. This could allow the use of cheaper oscillators with lower frequency stability in the receiver tuning circuits. 
     Second Bootstrap OFDM Symbol Processor 
     An example block diagram of a second bootstrap processor  1610  shown in  FIG. 16  will now be described in more detail with reference to  FIG. 22 .  FIG. 22  provides a schematic block diagram of an example embodiment to the present technique which is arranged to generate an estimate of the signalling information, which is represented by a cyclic shift which has been applied to the signature sequence at the transmitter. As shown in  FIG. 22  the received signal is fed on channel  1801  in the frequency domain from the output of the FFT processor  58  to a divider  2201 . A divider  2201  receives on a second input  2202  an estimate of the channel transfer function H(k), which has been calculated for example by the first bootstrap processor/decoder  1602  as explained above. The divider  2201  divides the received signal R(k) by the estimate of the channel transfer function H(k) which therefore equalises the received OFDM symbol. The received OFDM symbol after equalisation is fed to a second divider  2204  which receives on a second input  2206  the signature sequence C n (k) corresponding to the n-th OFDM bootstrap symbol, where for the present example n=2, 3 or 4. The equalised received signal is then divided by the signature sequence C n (k) fed from a data store  2208  through a channel  2206  and the output is fed to a slope estimator  2210 . The slope estimator estimates the average change of the phase from sub-carrier to sub-carrier. The slope estimator outputs a phase slope θ with units of radians per sub-carrier through channel  2212 . To estimate the shift which has been applied to the signature sequence to represent the signalling information as explained above, this phase slope first has to be converted into units of degree per Hertz. This is done by feeding θ from an output channel  2212  to a multiplier  2214  and multiplied with −F s /2πΔf where F s  is the sampling frequency of the bootstrap signal and Δf is the sub-carrier bandwidth used for each bootstrap OFDM symbol. In some embodiments, F s =6.144E6 and Δf=3000. The result of the multiplication is fed from an output of a multiplier  2220  as an estimate of the signalling information, because, as will be explained below, the phase slope represents the cyclic shift applied to the signature sequence at the transmitter. 
     The phase slope estimate θ from the slope estimator  2210  is then fed to a second multiplier  2230  which also receives an indication of the sub carrier count k. The multiplier  2230  multiplies the phase slope by the sub-carrier number k. The output of multiplier  2230  is a phase which is output into an exponential generator  2234  which uses the input phase to generate a sinusoid. This phase shift is generated for each of the sub-carriers k and fed to a first input  2236  of a third multiplier  2238 . On a second input  2240  the frequency domain OFDM symbol is received and multiplied by the phase shift thereby removing the effect of the detected cyclic shift of the signature sequence from the OFDM symbol. The output of the multiplier  2238  is fed to the first input of a divider  2242  which also receives the signature sequence for the n-th bootstrap OFDM symbol C n (k) from the store  2208  from the connecting channel  2206 . The output of the divider  2242  therefore also forms an estimate of the channel transfer function for the n-th bootstrap OFDM symbol H n (k). The estimate of the channel transfer function is then multiplied using a fourth multiplier  22  by a factor α from a factor unit  2252 . The output of the multiplier  2250  is then fed to a first input of an adder  2254  which receives via a second input  2256  an output from a fifth multiplier  2258 , which receives on a first input a factor 1−α  2260  and a second input a delayed value of the output of the adder  2264 . The output of the adder  2254  forms a combination of the channel transfer function for the current symbol n combined with an estimate of the channel transfer function from the previous symbol n−1 as delayed by a delay unit  2266 . Therefore, in combination for each of the received OFDM symbols of the bootstrap symbol, an update of the channel transfer function is performed based on an estimate of the channel transfer function for each successive bootstrap OFDM symbol. 
     Bootstrap Decoding in the Frequency Domain 
     According to the present technique an estimate of the cyclic shift of the signature sequence is generated by estimating the phase slope across sub-carriers in the frequency domain that results from the cyclic shift of the signature sequence in the time domain by the slope estimator  2210 . This arrangement is shown in isolation in  FIG. 23  in order to provide a more thorough explanation and so parts appearing in  FIG. 23  have the same numerical designations as in  FIG. 22 . 
     As explained above, the transmitter applies a cyclic shift of m-samples to the signature sequence, so that 
                 r   n     ⁡     (   i   )       ⁢     →   yields     ⁢         r   n     ⁡     (     i   -   m     )       .           
By the cyclic shift property of the FFT:
 
FFT( r   n ( i−m ))= e   −j2πmk/N   R   n ( k ), where  R   n ( k )=FFT( r   n ( i )
 
     The component e −j2πmk/N  can be used to estimate the cyclic shift representing the signal information {tilde over (M)} n  for the bootstrap symbols BS [n]  where n=2, 3, 4 Therefore by executing an FFT on the useful part of the bootstrap OFDM symbol r n (i) R n (k) is obtained. Then perform a zero-forcing equalisation on R n (k) using Ĥ n (k) to obtain R′ n (k) according to:
 
 R′   n ( k )= R   n ( k )÷ Ĥ   n ( k )
 
     As shown in  FIG. 23 , the FFT processor  58  feeds the frequency domain OFDM symbol to the first divider  2201  which equalises the received signal by dividing the received symbol in the frequency domain R(k) by the current estimate of the channel transfer function Ĥ n (k). A second divider  2202  then divides the equalised received symbol from the output of the first divider  2201  by the signature sequence for the n-th symbol C n (k) to isolate a frequency shift on each of the sub-carriers of the OFDM symbol which is produced as a result of a cyclic shift applied to the signature sequence at the transmitter. The effect of the signature sequence in the frequency domain is then removed to obtain R″ n (k) according to the expression:
 
 R″   n ( k )= R′   n ( k )÷ C   n ( k )
 
The phase component of R″ n (k) is then measured as φ(k)=Arg(R″ n (k)). This argument is calculated for k=(−N ZC −1)/2 to (N ZC −1)/2, so that
 
     
       
         
           
             θ 
             = 
             
               
                 2 
                 
                   
                     N 
                     ZC 
                   
                   - 
                   1 
                 
               
               ⁢ 
               
                 
                   ∑ 
                   
                     k 
                     = 
                     1 
                   
                   
                     k 
                     = 
                     
                       
                         ( 
                         
                           
                             N 
                             ZC 
                           
                           - 
                           1 
                         
                         ) 
                       
                       / 
                       2 
                     
                   
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     ( 
                     
                       
                         ϕ 
                         ⁡ 
                         
                           ( 
                           k 
                           ) 
                         
                       
                       - 
                       
                         ϕ 
                         ⁡ 
                         
                           ( 
                           
                             - 
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     As shown in  FIG. 23 , a phase slope estimator  2204  performs a function of unwrapping a phase curve caused by alternative signs on each carrier so that addition elements  2206 ,  2208  can detect a slope of the function line  2210  of phase  2212  against frequency  2214 . The phase slope estimator  2204  therefore measures the slope on Q(k)=Arg(R″ n (k)) across all subcarriers to determine θ rads per FFT bin. The output of the phase slope estimator  2204  then produces a phase value θ on a channel  2216  which is fed to a multiplier  2218 . 
     The cyclic shift is then determined as: 
                 M   ~     n     =       -     θ     2   ⁢   πΔ   ⁢           ⁢   f         *     F   S             
where Δf is the subcarrier bandwidth, for example 3000 Hz and F S  is the baseband sampling frequency of the bootstrap signal, for example 6.144 MHz. As such, the multiplier  2218  also receives a number representing
 
             (     -       F   S       2   ⁢   πΔ   ⁢           ⁢   f         )         
which when multiplied by the phase slope produces an indication of the cyclic shift and therefore the signalling information which was conveyed by the bootstrap OFDM symbol at the transmitter. Accordingly,  FIGS. 22 and 23  provide an example in which the bootstrap processor/decoder  1610  shown in  FIG. 16 , which detects the signalling information conveyed with the bootstrap symbol in the frequency domain.
 
     In another embodiment, as shown in  FIG. 24 , the output of the divider  2202  is transformed back into the time domain using an inverse FFT  2401  and the cyclic shift is detected by the peak detector  2402 , from the sample number of the peak sample in the transform result. The signalling information {tilde over (M)} n  is then detected by subtracting a total number of samples  2048  from sample number of the peak, using a subtraction circuit  2406 . According to the example embodiment shown in  FIG. 24 , a more robust technique for estimating the cyclic shift is provided, which does not require the estimation of the phase slope. The phase slope method can be highly sensitive to noise and also requires resolution of a phase slope ambiguity when the cyclic shift is exactly  1024 . 
     Channel Transfer Function Update 
     As explained above with reference to  FIG. 22  the bootstrap processor for detecting the signalling information according to the present technique includes an arrangement for tracking changes in the channel transfer function Ĥ n (k) between OFDM symbols of the bootstrap signal. In particular, elements which fall within the box  2280  shown in  FIG. 22  provide a “leaky bucket” arrangement for adapting the channel transfer function which is used to equalise the next OFDM symbol with respect to an estimated version of the channel transfer function for the current OFDM symbol. This is because the first estimate of the channel transfer function Ĥ 1 (k), which is generated from the first bootstrap OFDM symbol (BS 1 ) may have changed for subsequently transmitted OFDM symbols of the bootstrap signal (BS [n] , where n=2, 3 4), particularly for fast moving channels. According to the present technique therefore, the channel transfer function Ĥ 2 (k) which is used to equalise the n-th bootstrap symbol is updated for each bootstrap symbol. In particular, units α  2252  and 1−α  2261  have an effect of scaling each of the coefficients of the channel transfer function for the estimated version of the current symbol with respect to the estimated version for the previous symbol which are then combined by the added  2254  to form an estimate of the channel transfer function for the next OFDM symbol. As such following the delay  2266 , the estimated channel transfer function for the next OFDM symbol is derived from a combination of the channel transfer function for the current OFDM symbol and the estimated channel transfer function for the previous symbol. The following paragraphs provide a more detailed explanation of the operations of the bootstrap processor to update the channel transfer function and equalise the received signal: 
     First the cyclic shift on the n-th symbol is determined r n (i), and this is used to remove the effect of the cyclic shift to form r′ n (i) by reversing the shift. The reversal of the phase shift can be done in the time domain or in the frequency domain. For the time domain removal of the cyclic shift, the samples of the time domain symbol are simply shifted cyclically by a number of cycles equal to the cyclic shift to be removed. The adapted received symbol r′ n (i) is then transformed into the frequency domain according to the expression:
 
 R′   n ( k )=FFT( r′   n ( i ))
 
     For the frequency domain removal of the channel transfer function, the received bootstrap OFDM symbol is first transformed into the frequency domain according to the expression:
 
 R   n ( k )=FFT( r   n ( i ))
 
     Since the cyclic shift is estimated in the frequency domain as a phase slope θ, the phase slope can then be removed in the frequency domain according to:
 
 R′   n ( k )= R   n ( k ) e   jkθ 
 
     The channel transfer function for the currently received n-th OFDM symbol is then determined as
 
 H   n ( k )= R′   n ( k )/ C   n ( k ) where  n= 2,3,4
 
     Then the channel transfer function for the next bootstrap OFDM symbol (n+1) is determined according to:
 
 Ĥ   n+1 ( k )=α H   n ( k )+(1−α) H   n−1 ( k )
 
     Therefore following a delay of one symbol the updated channel transfer function is used for the current symbol as Ĥ n (k). The factor α can be decided experimentally, for example ⅔. 
     Bootstrap Decoding in the Time Domain 
     In contrast to  FIGS. 22, 23 and 24  which show the bootstrap decoder operating to detect the signalling information from the shift in the signature sequence in the frequency domain,  FIG. 15  provides an example in which the bootstrap decoding is performed in the time domain. That is the shift of the signature sequence carried by the OFDM symbol is detected in the time domain. According to the arrangement shown in  FIG. 15  the look up table/signature sequence generator  1534  provides the signature sequence on channel  1536  to the multiplier  1530  which is multiplied by the estimate of the channel transfer function Ĥ n (k) by the multiplier  1530 . This forms effectively a channel shaping of the signature sequence. The shaped signature sequence is then transformed from the frequency domain to the time domain as it appears at the output of the Inverse Fast Fourier Transform (IFFT) processor  1540  on channel  1542 . The isolated useful part of the received OFDM symbol is then fed from channel  1522  to a relative cyclic shift estimation circuit. The relative cyclic shift estimation circuit  1524  performs a time domain correlation of the received signal with the channel shaped time domain signature sequence in order to detect the cyclic shift of the signature sequence and therefore the signalling data M n . This operation is expressed mathematically as follows: 
     According to the present technique the time domain estimate of the relative shift of the signature sequence {tilde over (c)} n (i) is determined for the n-th bootstrap symbol by first channel shaping the signature sequence, by multiplying the sequence in the frequency domain by the estimate of the channel transfer function for the n-th symbol according to:
 
 {tilde over (C)}   n ( k )= C   n ( k ) Ĥ   n ( k )
 
     Then the resulting sequence is transformed into the time domain by performing an inverse Fourier Transform according to:
 
 {tilde over (c)}   n ( i )=IFFT( {tilde over (C)}   n ( k ))
 
     The useful part of the received signal is then isolated by the B−C remover  1520  so that the part r n (i) for the n-th bootstrap OFDM symbol is then correlated with the channel shaped time domain signature sequence according to the expression:
 
 g   n ( l )=Σ i=0   i−N     FFT     −1   {tilde over (c)}   n ( i ) r   n (mod( i+ 1, N   FFT )) for  l= 0,1, . . . , N   FFT  
 
     As a result of the structure of the signature sequence the correlation is circular so that r n (i)=r n (i mod N FFT ), where for example N FFT =2048. Accordingly, the correlation requires N FFT *N FFT  complex multiplies and N FFT *N FFT  complex adds. The relative cyclic shift is then determined in the time domain as: 
     
       
         
           
             
               
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     The absolute cyclic shift is then decoded as:
 
 M   n =mod( M   n−1   −{tilde over (M)}   n   ,N   FFT ), with  M   1 =0
 
     According to an example embodiment of the present technique the relative cyclic shift maybe estimated in the time domain by using a technique, which decimates the signature sequence in order to reduce the complexity of the cyclic shift estimator. 
     According to the present technique the samples of the received signal and the channel shaped signature sequence in the time domain can be decimated in order to reduce a complexity with which the estimation of the cyclic shift is determined.  FIG. 25  provides an example illustration of a decimation performed to generate the cyclic shift estimation in the time domain. As shown in  FIG. 25  a first line of cells represents 21 samples of the channel shaped signature sequence in the time domain whereas line  2502  represents the 21 samples of the received signal in the time domain. For the first lines  2501 ,  2502  a cyclic shift estimation maybe formed by multiplying each of the samples together in order to determine the cyclic shift as a cross correlation sample. However, it is possible to estimate the cyclic shift by decimating both the samples of the channel shaped signature sequence and the received signal  2501 ,  2502 . Third and fourth lines shown in  FIG. 2504, 2506  show some elements, which are shaded and some which are clear. The elements which are shaded are used in the multiplication to perform the cross correlation to estimate the cyclic shift and those with a clear or white filling are not used. Accordingly, the lines  2504 ,  2506  represent a decimation of d=2 that is, every other sample is used in the generation of the cyclic shift estimation. Fifth and sixth lines  2508 ,  2510  provide a further example in which the decimation value d=4. That is to say, only one in four of the samples of the signature sequence  2508  and the received signal  2510  are used to calculate the cross correlation between the channel shaped signature sequence and the received signal in the time domain and therefore estimate the signalling information carried by the bootstrap OFDM symbol. Accordingly, only one in four of the cells in each of the received signal and the channel shaped signature sequence are shaded. This decimation and simplification can be expressed according to the above equations as follows:
 
 g   n ( l )=Σ i=0,i+=D   i−N     FFT     −1   {tilde over (c)}   n ( i ) r   n (mod( i+ 1, N   FFT ))
 
     According to this arrangement, the correlation requires N FFT *N FFT ÷D complex multiplies and N FFT *N FFT ÷D complex adds. As such for an example in which the decimation factor D=8, then each iteration l now requires only 
                 N   FFT     8     =   256         
multiply and add operations.
 
     According to an example embodiment of the present technique therefore a receiver for detecting and recovering payload data from a received signal comprises a radio frequency demodulation circuit configured to detect the received signal. A bootstrap processor is configured to detect the signalling data from the bootstrap OFDM symbols using an estimate of the channel transfer function determined from one or more of the bootstrap OFDM symbols, and a demodulator circuit is configured to recover the payload data from the payload OFDM symbols using the signalling data. The bootstrap processor may include a signature sequence generator, which is configured to regenerate a copy of the signature sequence. The bootstrap processor comprises a channel shaper, which is configured to convolve the signature sequence with the estimate of the channel transfer function to generate a channel shaped signature sequence, a cross-correlator and a cyclic shift detector. A cross-correlator is configured to cross-correlate the useful part of each of the one or more bootstrap OFDM symbols with the regenerated and channel shaped copy of the signature sequence, and the cyclic shift detector is configured to estimate the signalling data conveyed by each of the one or more bootstrap OFDM symbols by detecting a cyclic shift of the signature sequence present in each of the one or more bootstrap OFDM symbols from a peak of the samples representing a result of the cross-correlation. 
     According to the present technique therefore, determining the cyclic shift of the signature sequence carried by the received OFDM symbols of the bootstrap signal in the time domain can be provided with a reduced complexity and therefore an improved performance of the receiver. For example decimation techniques can be used to reduce the complexity of calculating the cross-correlation. 
     Summary of Operation 
     In summary a receiver configured to detect a bootstrap signal in accordance with the present technique performs the operations as represented in  FIGS. 26 and 27 .  FIG. 26  represents the operation of the bootstrap processor for the first and subsequent bootstrap symbols whereas  FIG. 27  represents the operations performed for the first bootstrap processor for detecting the signature sequence, integer and fine frequency offsets and the initial channel transfer function. 
     The following diagram shown in  FIG. 26  is therefore summarised as follows: 
     S 1 : At the start of the process the received symbol is in the frequency domain as shown in  FIG. 16 . In some embodiments, the symbol spectrum may be oversampled having been derived from an FFT of larger than 2048 spectral output components. 
     S 2 : At the start of the loop the reference signature sequence index variable is initialised as i=1. 
     S 4 : A cyclic correlation is therefore performed between the received frequency domain OFDM symbol and the i-th signature sequence. 
     S 6  and S 8 : A significant peak value will be detected in the output of the cross correlation within the range of IFO if the reference signature sequence i is the same as the one used at the transmitter. If significant peak value has not been detected then processing proceeds to step S 10  and the reference signature sequence index variable i increased and the next reference signature sequence is tried in the cross-correlation. The candidate reference signature sequences may be pre-stored in the receiver with the indexing based a combination of the root of the Zadoff-Chu and the seed of the PN generator used to generate the particular sequence.
 
S 12 : If a significant peak of the cross-correlation has been detected, then the current value of i is the index to the wanted reference signature sequence; the relative position of the peak in the cross-correlation output is used to determine the Integer Frequency Offset (IFO) and the Fine Frequency Offset (FFO) if the spectrum was over-sampled.
 
S 14 : At step S 14  the received signal is divided by the detected signature sequence to generate an estimate of the channel transfer function H(k). At step S 16  the process ends.
 
     The flow diagram shown in  FIG. 27  which represents the operation for all the bootstrap symbols is summarised as follows: 
     S 20 : At the start of the process the value n for the bootstrap symbol which can be two, three or four is set to one. 
     S 22 , S 23 : CAB correlation is used to detect the presence of the first bootstrap symbol. The CAB correlator is run continuously until a significant peak is detected at its output (S 23 ). The occurrence of the peak is used to trigger the FFT that converts this first OFDM symbol into the frequency domain (S 24 ) whilst the argument of the peak sample is used to estimate the fine frequency offset (FFO) in step S 26 .
 
S 28 : This step is the process described above in  FIG. 25 . The Integer Frequency Offset (IFO) estimation, and the signature sequence index is then determined. If spectrum over-sampling was used, a more refined FFO may also be estimated.
 
S 30 : With knowledge of both the IFO and the FFO, the combined frequency offset is then removed from the received signal and the channel transfer function estimated in step S 32  by dividing the received signal from which the frequency offset has been removed by the reference signature sequence identified in step S 28 .
 
     Processing then proceeds with an increase in the variable n because the next bootstrap symbol is being processed. At step  34  therefore the value of n is incremented. The next steps depend on whether the receiver is designed to detect the cyclic shifts of the signature sequence in the subsequently received bootstrap symbols in the time domain or the frequency domain. If in the time domain, then processing will proceed to S 38  otherwise, the next step is S 44   
     S 38 : In the time domain the signature sequence is first shaped by the channel transfer function estimated in step S 32  and then converted into the time domain using an Inverse Fast Fourier Transform performed at step S 40 . At step S 42  a cross correlation of the channel shaped time domain signature sequence is performed with the received symbol in order to detect the cyclic shift and therefore the signalling information carried by the OFDM symbol.
 
S 44 : In contrast if processing is performed in the frequency domain then the received signal is transformed into the frequency domain at step S 44  and at step S 46  the frequency domain symbol is equalised by dividing the symbol by the estimate of the channel transfer function.
 
S 48 : The equalised signal is then divided by the frequency domain version of the signature sequence and an estimate is performed of a slope of the resulting phase shift across the OFDM sub-carriers at step S 50 . At step S 52  the slope of the change of phase with respect to sub-carrier is then determined and used to calculate the cyclic shift representing the information carried by the bootstrap symbol. At step S 54  it is determined whether the last OFDM symbol has been received. If it is the last OFDM symbol of the bootstrap signal then processing stops at step S 56  otherwise the channel transfer function is updated in step S 56  as explained above and processing proceeds again to determine the cyclic shift of the signature sequence in the next bootstrap symbol in order to detect and decode the signalling information.
 
     Accordingly embodiments of the present technique can provide an arrangement for improving an accuracy with which the timing of the FFT trigger point for capturing a useful part of the OFDM symbol of the bootstrap signal is estimated. Correspondingly an estimate of the FFO is also improved. This is achieved by arranging for each of a plurality of correlators each of which is adapted to match the time domain structure of the different types of bootstrap OFDM symbols. By respectively delaying each of the correlation results from the respective correlators in accordance with a relative delay in transmission of a useful section of each of the OFDM symbols, the correlation results can be combined to produce a more accurate estimate of the FFT trigger point and the FFO. 
     The following numbered paragraphs define further example aspects and features of the present technique: 
     Paragraph 1. A receiver for detecting and recovering payload data from a received signal, the receiver comprising 
     a radio frequency demodulation circuit configured to detect and to recover the received signal, the received signal having been formed and transmitted by a transmitter to carry the payload data in Orthogonal Frequency Division Multiplexed (OFDM) symbols in one or more of a plurality of time divided frames, each frame including a preamble including a plurality of bootstrap OFDM symbols, one or more of the bootstrap OFDM symbols of the preamble carrying signalling data represented as a relative cyclic shift of a signature sequence which has been combined with the one or more of the bootstrap OFDM symbols, 
     a bootstrap processor configured to detect the signalling data from the bootstrap OFDM symbols using an estimate of the channel transfer function determined from one or more of the bootstrap OFDM symbols, and 
     a demodulator circuit configured to recover the payload data from the payload OFDM symbols using the signalling data, wherein the bootstrap processor comprises 
     a channel shaper, which is configured to convolve a time domain copy of the signature sequence with the estimate of the channel impulse response to generate a channel shaped signature sequence, 
     a cross-correlator configured to cross-correlate the useful part of each of the one or more bootstrap OFDM symbols with the channel shaped time domain copy of the signature sequence, and 
     a cyclic shift detector configured to estimate the signalling data conveyed by each of the one or more bootstrap OFDM symbols by detecting a cyclic shift of the signature sequence present in each of the one or more bootstrap OFDM symbols from a peak of the samples representing a result of the cross-correlation. 
     Paragraph 2. A receiver according to paragraph 1, wherein the signature sequence which is combined with the one or more bootstrap OFDM symbols to represent the signalling data comprises a plurality of coefficients, and each of the coefficients of the signature sequence has mapped to one of the sub-carriers of the OFDM symbol followed by conversion to the time domain and cyclic shifting the time domain symbol to an amount which represents signalling information to be transmitted via the bootstrap symbol, and the detection circuit comprise 
     a channel estimator, which estimates a coefficient of the channel transfer function at each subcarrier, wherein the channel shaper is configured to convolve the generated signature sequence in the time domain with the channel impulse 
     Paragraph 3. A receiver according to paragraph 2, wherein the channel shaper is configured to multiply each of the coefficients of the signature sequence with a corresponding coefficient of the channel transfer function, and the channel shaper includes an inverse Fourier transform processor configured to convert the signature sequence multiplied with the channel transfer function from the frequency domain into the time domain, for cross-correlation with the time domain one or more bootstrap OFDM symbols carrying the signalling data by the cross-correlator. 
     Paragraph 4. A receiver according to paragraph 3, wherein a first of the plurality of bootstrap OFDM symbols has been combined with the signature sequence without a cyclic shift, and the bootstrap processor includes a channel estimator configured 
     to receive the first bootstrap OFDM symbol, 
     to convert a useful part of the first bootstrap OFDM symbol into the frequency domain, 
     to divide each of the sub-carriers of the frequency domain version of the first bootstrap OFDM symbol with a corresponding coefficient of the signature sequence, to generate for each of the coefficients of the signature sequence a corresponding coefficient of the channel transfer function. 
     Paragraph 5. A receiver according to any of paragraphs 1 to 4, wherein the cross-correlator is configured to decimate the signature sequence to select one from every d samples of the coefficients of the channel shaped signature sequence for generating the correlation result samples. 
     Paragraph 6. A receiver according to any of paragraphs 1 to 5, wherein the signature sequence comprises a combination of a Zadoff-chu sequence and a pseudorandom-noise sequence. 
     Paragraph 7. A method of detecting and recovering payload data from a received signal, the method comprising 
     detecting the received signal, the received signal having been formed and transmitted by a transmitter to carry the payload data in Orthogonal Frequency Division Multiplexed (OFDM) symbols in one or more of a plurality of time divided frames, each frame including a preamble including a plurality of bootstrap OFDM symbols, one or more of the bootstrap OFDM symbols of the preamble carrying signalling data represented as a relative cyclic shift of a signature sequence which has been combined with the one or more of the bootstrap OFDM symbols, 
     detecting the signalling data from the bootstrap OFDM symbols using an estimate of the channel transfer function determined from one or more of the bootstrap OFDM symbols, and 
     recovering the payload data from the payload OFDM symbols using the signalling data, wherein the detecting the signalling data from the bootstrap OFDM symbols comprises 
     convolving a time domain copy of the signature sequence with the estimate of the channel impulse response to generate a channel shaped signature sequence, 
     cross-correlating the useful part of each of the one or more bootstrap OFDM symbols with the channel shaped signature sequence, and 
     estimating the signalling data conveyed by each of the one or more bootstrap OFDM symbols by detecting a cyclic shift of the signature sequence present in each of the one or more bootstrap OFDM symbols from a peak of the samples representing a result of the cross-correlation. 
     Paragraph 8. A method according to paragraph 7, wherein the signature sequence which is combined with the one or more bootstrap OFDM symbols to represent the signalling data comprises a plurality of coefficients, and each of the coefficients of the signature sequence has been combined with one of the sub-carrier of the OFDM symbol followed by conversion to the time domain and cyclic shifting the time domain signal to an amount which represents signalling information to be transmitted via the bootstrap symbol, and the method includes estimating a coefficient of the channel transfer function at each subcarrier of the one or more bootstrap OFDM symbols carrying the signalling data. 
     Paragraph 9. A method according to paragraph 8, wherein the convolving of the signature sequence with the estimate of the channel impulse response comprises 
     multiplying each of the frequency domain coefficients of the signature sequence with a corresponding coefficient of the channel transfer function, and 
     performing an inverse Fourier transform to convert the signature sequence multiplied with the channel transfer function from the frequency domain into the time domain, for cross-correlation with the time domain one of more bootstrap OFDM symbols carrying the signalling data. 
     Paragraph 10. A method according to paragraph 9, wherein a first of the plurality of bootstrap OFDM symbols has been combined with the signature sequence without a cyclic shift, and the method includes 
     receiving the first bootstrap OFDM symbol, 
     converting a useful part of the first bootstrap OFDM symbol into the frequency domain, 
     dividing each of the sub-carriers of the frequency domain version of the first bootstrap OFDM symbol with a corresponding coefficient of the signature sequence, to generate for each of the coefficients of the signature sequence a corresponding coefficient of the channel transfer function. 
     Paragraph 11. A method according to any of paragraphs 7 to 10, wherein the cross-correlator is configured to decimate the signature sequence to select one from every d samples of the coefficients of the signature sequence for generating the correlation result samples. 
     Paragraph 12. A method according to any of paragraphs 7 to 11, wherein the signature sequence comprises a combination of a Zadoff-chu sequence and a pseudorandom-noise sequence. 
     Various further aspects and features of the present technique are defined in the appended claims and various combinations of the features of the dependent claims may be made with those of the independent claims other than the specific combinations recited for the claim dependency. Modifications may also be made to the embodiments hereinbefore described without departing from the scope of the present technique. For instance, processing elements of embodiments may be implemented in hardware, software, and logical or analogue circuitry. Furthermore, although a feature may appear to be described in connection with particular embodiments, one skilled in the art would recognise that various features of the described embodiments may be combined in accordance with the present technique.
     [1] ATSC Candidate Standard: System Discovery and Signaling (Doc. A/321 Part 1), Document S32-231r4, 6 May 2015   [2] EN 302 755 V1.3.1, Frame structure channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), April 2012