Patent Publication Number: US-11394326-B2

Title: Control method and control device for electric vehicle

Description:
TECHNICAL FIELD 
     The present invention relates to a control method and a control device for an electric vehicle. 
     BACKGROUND ART 
     Conventionally, an oscillation suppression that reduces the torsional oscillation of a drive shaft connecting a motor to drive wheels is used as a control method for an electric vehicle powered by a synchronous motor using a permanent magnet for a rotor. 
     SUMMARY OF INVENTION 
     However, the magnetic flux generated in the motor fluctuates in the motor which does not have the permanent magnet, in contrast to the above-mentioned synchronous motor where the rotor magnetic flux is constant. Therefore, it is difficult to apply the above-mentioned oscillation suppression control without modification to the field magnet winding type synchronous motor that does not have the permanent magnet in the rotor. 
     On the other hand, JP5939316B discloses a method of applying the above-mentioned oscillation suppression control to an induction motor where a rotor flux fluctuates. However, since JP5939316B discloses the control method that applies the above-mentioned suppression control to the induction motor by compensating the torque current based on the excitation current (y-axis current). Therefore, because d-axis current and the current flowing in the field winding of the rotor (f-axis current) are necessary to be taken into account, it is not possible to apply this control method to the field magnet winding type synchronous motor. 
     The object of the present invention is to provide a technique for applying oscillation suppression control to the field magnet winding type synchronous motor in order to reduce torsional oscillation of the drive shaft connecting the motor to the drive wheels. 
     According to an aspect of the present invention disclosure, a method for controlling an electric vehicle having a field winding type synchronous motor for providing a driving force, the field winding type synchronous motor that has a rotor having a rotor winding and a stator having a stator winding, by controlling a stator current flowing in the stator winding and a rotor current flowing in the rotor winding is provided. The method includes: setting a basic torque command value based on a vehicle information; calculating a d-axis current command value and a first q-axis current command value for the stator current, and a f-axis current command value for the rotor current, based on the basic torque command value and the vehicle information; calculating a magnetic flux estimate value, which is an estimated value of a magnetic flux generated in the rotor, based on the d-axis current command value and the f-axis current command value; calculating a final torque command value, based on the first q-axis current command value and the magnetic flux estimate value; calculating a second q-axis current command value, based on the magnetic flux estimate value and the final torque command value. The method includes further includes controlling the stator current and the rotor current, based on the second q-axis current command value, the d-axis current command value and the f-axis current command value. 
     The embodiments of the present invention will be described in detail below referring to the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  illustrates a schematic diagram of the vehicle system to which a control method of an electric vehicle of a first embodiment illustrates applied. 
         FIG. 2  illustrates a flowchart showing a flow of processing performed by an electric motor controller. 
         FIG. 3  illustrates a figure showing an example of an accelerator opening-torque table. 
         FIG. 4  illustrates a block diagram of the motor control system of the first embodiment. 
         FIG. 5  illustrates a block diagram of a q-axis current control unit. 
         FIG. 6  illustrates a block diagram of a d-axis current control unit. 
         FIG. 7  illustrates a block diagram of a f-axis current control unit. 
         FIG. 8  illustrates a block diagram of an oscillation suppression control processing unit. 
         FIG. 9  illustrates a block diagram of a magnetic flux estimate unit. 
         FIG. 10  illustrates a block diagram of a reluctance-torque equivalent flux estimator of the first embodiment. 
         FIG. 11  illustrates a block diagram of a reluctance-torque equivalent flux estimator of the first embodiment. 
         FIG. 12  illustrates a figure illustrating the equation of motion of the electric vehicle. 
         FIG. 13  illustrates a time chart showing control results by the control method of the electric vehicle of the first embodiment. 
         FIG. 14  illustrates a block diagram of a motor control system of a second embodiment. 
         FIG. 15  illustrates a block diagram of a f-axis current control unit. 
         FIG. 16  illustrates a block diagram of a f-axis F/F compensator. 
         FIG. 17  illustrates a block diagram of a f-axis current model. 
         FIG. 18  illustrates a block diagram of a f-axis current F/B model. 
         FIG. 19  illustrates a block diagram of a f-axis limit processing unit. 
         FIG. 20  illustrates another block diagram of the f-axis limit processing unit. 
         FIG. 21  illustrates a block diagram of the f-axis F/B compensator. 
         FIG. 22  illustrates a block diagram of the f-axis robust compensator. 
         FIG. 23  illustrates a flowchart showing the motor control processing. 
         FIG. 24  illustrates a block diagram of a magnetic flux estimator of the second embodiment. 
         FIG. 25  illustrates a block diagram of a reluctance-torque equivalent flux estimator of the second embodiment. 
         FIG. 26  illustrates a time chart showing control results by the control method for the electric vehicle of the second embodiment. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     The First Embodiment 
       FIG. 1  is the block diagram showing an example of the configuration of the motor control system  100  to which the control method for the electric vehicle according to one of the embodiments of the present invention is applied. The electric vehicle is a type of a vehicle that is equipped with at least one field magnet winding type synchronous motor (hereinafter simply referred to as “the motor”) as part or all of the driving source of the vehicle and can run by the driving power of the motor, which includes electric vehicles and hybrid vehicles. 
     A battery  1  discharges the drive power from a field magnet winding type synchronous motor  4  and charges the regenerative power to and for the motor  4 . 
     A electric motor controller  2  (hereinafter referred to simply as “the controller”) includes, for example, a central processing unit (CPU), a read-only memory (ROM), a random-access memory (RAM), and an input/output interface (I/O interface). Signals of various vehicle variable parameters indicating the vehicle status, such as a vehicle speed V, an accelerator opening e, an electric angle θ re  of the motor  4 , stator currents of the motor  4  (i u , i v , and i w  in the case of three-phase AC), a rotor current (i f ) of the motor  4 , and the like, are inputted to the controller  2  as digital signals. The controller  2  generates PWM signals in order to control the motor  4  based on the inputted signals. The controller  2  further generates drive signals for the inverter  3  after receiving the generated PWM signals. 
     The inverter  3  converts the DC current supplied from the battery  1  to the AC current or oppositely inverts the latter to the former by turning on and off two switching elements (e.g., power semiconductor elements such as IGBT and MOS-FET) provided for each phase in order to control the stator current, such that the desired current flows in the motor  4 . With the inverter  3 , the two pairs (four in total) of the switching elements (e.g., the power semiconductor elements such as IGBT and MOS-FET) are connected to both ends of the rotor winding, in order to control the rotor current. By turning the switching elements on and off in response to the drive signal, the desired current flows through the rotor winding. However, if the number of directions of the current flowing in the rotor is only one, the two diagonally positioned switching elements of the two pairs of the switching elements may be replaced with diodes. 
     The field magnet winding type synchronous motor  4  (hereinafter simply referred to as “the motor  4 ”) is a field magnet winding type synchronous motor which includes the rotor with a rotor winding (field winding) and the stator with a stator winding (armature winding). When the motor control system  100  of this embodiment is mounted in the vehicle, the motor  4  works as the drive source for the vehicle. As will be described in detail later, the motor  4  is controlled by controlling the rotor current flowing through the rotor winding and the stator current flowing through the stator winding. The motor  4  generates a drive torque by the current supplied from the inverter  3 , and transmits the drive power to the right and left drive wheels  9  via the reduction gear  5  and a drive shaft  8 . In addition, the motor  4  collects the kinetic energy of the vehicle as electric energy by making a regenerating breaking force when the motor is rotated by the drive wheels  9  together with the motor while the vehicle is running. In this case, the inverter  3  inverts the AC current generated during the regeneration operation of the motor  4  into DC current and supplies the DC current to the battery  1 . 
     A current sensor  7  detects the three-phase currents i u , i v , and i w  (stator currents) flowing in the stator winding of the motor  4 , and also detects the current i f  (rotor current) flowing in the rotor winding of the motor  4 . However, for the stator current, since the sum of the three-phase AC currents i u , i v , and i w  is zero, the current of any two phases may be detected and the current of the remaining one phase may be obtained by calculation. 
     The rotation sensor  6  is, for example, a resolver or encoder that detects the rotor phase a of the motor  4 . 
       FIG. 2  illustrates a flowchart showing a flow of processing performed by the controller  2 . The processing from step S 201  to step S 204  is programmed in the controller  2  to be constantly executed at certain intervals while the vehicle system is running. 
     In step S 201 , a signal indicating the vehicle state is inputted to the controller  2 . Here, a vehicle speed V (km/h), an accelerator opening θ (%), an electric angle θ re  of the motor  4 , a motor rotation speed Nm (rpm) of the motor  4 , the currents i u , i v , i w , and i f  flowing in the motor  4 , and a DC voltage V dc  (V) of the battery  1  are inputted to the controller  2 . 
     The vehicle speed V (km/h) is obtained by the communication from a meter (not shown), while examples of the meter are a vehicle speed sensor, or another controller such as a brake controller. Alternatively, the controller  2  obtains the vehicle speed V (m/s) by multiplying the rotor mechanical angular velocity ω m  by a tire dynamic radius r, and dividing the product of the multiplication by the gear ratio of the final gear, and then multiplying the quotient of the division by the unit conversion factor from m/s to km/s (3600/1000). 
     The accelerator opening θ (%) is obtained from the accelerator opening sensor, which is not shown in the figure. The accelerator opening θ (%) may be made to be obtained from other controllers such as a vehicle controller not shown in the figure. 
     The electric angle θ re  (rad) of the motor  4  is obtained from the rotation sensor  6 . The rotation speed N m  (rpm) of the motor  4  is obtained by dividing the electric angular velocity co, by the number of pole pairs p of the electric motor to obtain the motor rotation speed detection value ω m  (rad/s), which is the mechanical angular velocity of the motor  4 , and then multiplying the obtained motor rotation speed detection value ω m  by the unit conversion factor (60/(2π)) from rad/s to rpm. 
     The currents i u , i v , i w , and i f  (A) flowing in the motor  4  are obtained by the current sensor  7 . 
     The DC voltage V dc  (V) is detected by the voltage sensor (not shown) mounted in the DC power line between the battery  1  and the inverter  3 . The DC voltage V dc  (V) may be detected by a signal transmitted from a battery controller (not shown). 
     In step S 202 , a motor torque command value calculation process is executed. In the motor torque command value calculation process, a motor torque command value (basic torque command value) T m * is set by referring to an accelerator opening-torque table shown in  FIG. 3  based on the accelerator opening e and vehicle speed V inputted in step S 201 . 
     In step S 203 , an oscillation suppression control calculation process is executed. Specifically, the controller  2  calculates a q-axis current command value i q2 *, a d-axis current command value i d1 *, and a f-axis current command value i f1 * that suppress the driving force transmission system oscillation (torsional oscillation of the drive shaft  8 , etc.) without losing the response of the drive shaft torque based on the motor torque command value T m * set in step S 202 . The details of the oscillation suppression control calculation process will be described later. 
     In step S 204 , a current control calculation processing is performed. In the current control calculation process, a current control is performed to make a d-axis current i d , a q-axis current i q , and a f-axis current i f  match the q-axis current command value i q2 *, the d-axis current command value i d1 * and the f-axis current command value i f1 * calculated in step S 203 , respectively. The details of the current control calculation process will be explained using  FIG. 4  below. 
       FIG. 4  illustrates a figure showing an example of the configuration of the motor control system  100  and alternatively a control block diagram of a current control calculation processing unit  2   a , which is provided in the controller  2  as a functional section. The controller  2  executes the current control calculation processing shown in step S 204  using the current control calculation processing unit  2   a.    
     The current control calculation processing unit  2   a  includes a stator PWM conversion unit  401 , a rotor PWM conversion unit  402 , a look-ahead compensation unit  403 , coordinate conversion units  404 ,  410 , a non-interference control unit  405 , a q-axis current control unit  406 , a d-axis current control unit  407 , an f-axis current control unit  408 , a voltage command value calculation unit  409 , and an A/D converter  411 . 
     The stator PWM conversion unit  401  generates PWM_Duty drive signals (power electric element drive signals) D uu *, D ul *, D vu *, D vl , D wu *, D wl * provided in the inverter  3  and outputs the signals to the inverter  3  based on three-phase voltage command values v u *, v v *, and v w * outputted from a coordinate conversion unit  410  described below. 
     The rotor PWM conversion unit  402  generates PWM_Duty drive signals D fu * and D fl * for the rotor switching elements provided in the inverter  3  based on a f-axis voltage command value v f * described below and outputs the generated signals to the inverter  3 . 
     The inverter  3  generates the AC voltages v u , v v , and v w  for controlling the d-axis currents i d  and the q-axis current i q  flowing in the rotor winding of the motor  4  based on the PWM_Duty drive signals generated in the stator PWM conversion unit  401  and supplies the signals to the motor  4 . In addition, the inverter  3  generates the f-axis voltage v f  for controlling the f-axis current i f  flowing in the rotor winding of the motor  4  based on the PWM_Duty drive signal generated by the rotor PWM converter  402 , and supplies the signal to the motor  4 . 
     The current sensor  7  detects currents in at least two phases, e.g. the u-phase current i u  and v-phase current i v , of the three-phase AC current supplied from the inverter  3  to the motor  4 . The detected two phase currents i u  and i v  are converted into digital signals by the A/D (analog/digital) conversion unit  410  and are inputted to the coordinate transformation unit  404 . The current sensor  7  simultaneously detects the f-axis current i f  supplied to the motor  4  from the inverter  3 . The detected f-axis current i f  is converted into a digital signal by the A/D converter  411 , and is outputted to the f-axis current control unit  408 . 
     The look-ahead compensation unit  403  accepts the electric angle θ re  and the electric angular velocity ω re , and calculates a look-ahead compensated electric angle θ re ′ by adding a multiplication product to the electric angle θ re , while the multiplication product is calculated by multiplying the electric angular velocity ω re  and the latent time that the control system has missed. The look-ahead compensated electric angle θ re ′ is outputted to the coordinate transformation unit  404 . 
     The coordinate transformation unit  404  performs a transformation step from the three-phase AC coordinate system (uvw axes) to the orthogonal two-axes DC coordinate system (d-q axes). Specifically, the coordinate transformation unit  404  calculates the d-axis current i d  and the q-axis current i q  by performing the coordinate transformation process from the u-phase current i u , the v-phase current i v , the w-phase current i w , and the electric angle θ re , using the following equation (1). 
     
       
         
           
             
               
                 
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     In a non-interference control unit  405  accepts a d-axis current normative response i d_ref , a q-axis current normative response i q_ref , a f-axis current normative response i f_ref , a d-axis current normative response derivative value, s·i d_ref , and a f-axis current normative response derivative value s·i f_ref , and calculates non-interference voltages v d_dcpl  and v d_dcpl  and v f_dcpl  for offsetting the interference voltages between the d-axis, q-axis, and f-axis using the voltage equation shown by the equation (2) below. The equation (2) below is a voltage equation of the field magnet winding type synchronous motor  4 , which is the control target of the present embodiment. 
     
       
         
           
             
               
                 
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     Where each parameters of the above equation (2) represent as follows. Note that “s” in the equation is Laplace operation indicator. 
     I d : d-axis current 
     I q : q-axis current 
     I f : f-axis current 
     V d : d-axis voltage 
     V q : q-axis voltage 
     V f : f-axis voltage 
     L d : d-axis inductance 
     L w : q-axis inductance 
     L f : f-axis inductance 
     M: Mutual inductance between the stator and rotor 
     L d ′: d-axis dynamic inductance 
     L q ′: q-axis dynamic inductance 
     L f ′: f-axis dynamic inductance 
     M′: Dynamic mutual inductance between the stator and rotor 
     R a : Stator winding resistance 
     R f : Rotor winding resistance 
     ω re : Electric angular velocity 
     If the non-interference control by the non-interference control unit  405  functions ideally, the voltage equation in the equation (2) above can be diagonalized as shown in the equation (3) below. 
     
       
         
           
             
               
                 
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     According to the above equation (3), the characteristic values for the d-axis, q-axis, and f-axis from the voltage to the current become the primary delay as shown in the equations (4), (5), and (6) below, respectively. 
     
       
         
           
             
               
                 
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     The q-axis current control unit  406  calculates the first q-axis voltage command value v q_dsh , such that the measured value of the actual current (real current) is followed by the q-axis current command value i q2 * (the second q-axis current command value) with the desired response having no steady-state deviation, and outputs the first q-axis voltage command value v q_dsh  to the voltage command value calculation unit  409 . The details of the q-axis current control unit  406  will be described later using  FIG. 5 . 
     The d-axis current control unit  407  calculates the first d-axis voltage command value v d_dsh , such that the measured value of the actual current (real current) is followed by the d-axis current command value i d1 * with the desired response having no steady-state deviation, and outputs the first d-axis voltage command value v d_dsh  to the voltage command value calculation unit  409 . The details of the d-axis current control unit  407  will be described later using  FIG. 6 . 
     The f-axis current control unit  408  calculates the first f-axis voltage command value v f_dsh , such that the measured value of the actual current (real current) is followed by the f-axis current command value i n * with the desired response having no steady-state deviation, and outputs the first f-axis voltage command value v f_dsh  to the voltage command value calculation unit  409 . The details of the f-axis current control unit  408  will be described later using  FIG. 7 . 
       FIG. 5  illustrates a diagram showing the detail of the q-axis current control unit  406  of the present embodiment. The q-axis current control unit  406  includes a control block  501 , gain blocks  502 ,  503 , an integrator  504 , a subtractor  505 , and an adder  506 . 
     The control block  501  represents a transmission characteristic 1/(τ q s+1) of the primary delay that simulates the response delay of the actual current i q2 * for the q-axis current command value i q2 *. The control block  501  takes the q-axis current command value i q2 * as the input and outputs the q-axis current normative response i q_ref . Here, “τ q ” in the transmission characteristic 1/(τ q s+1) represents a q-axis current normative response time constant. 
     The gain block  502  represents a proportional gain K pq , which is expressed by the following equation (7). The gain block  502  takes as the input a deviation between the q-axis current command value i q2 * and the q-axis current i q , and outputs a value obtained by multiplying the input value by the proportional gain K pq  to the adder  506 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     7 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     
                       p 
                       ⁢ 
                       q 
                     
                   
                   = 
                   
                     
                       L 
                       q 
                       ′ 
                     
                     
                       τ 
                       q 
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
     The gain block  503  represents the proportional gain K iq , and is expressed by the following equation (8). The gain block  503  takes a deviation between the q-axis current command value i q2 * and the q-axis current i q  as the input, and outputs a value obtained by multiplying the input value by the proportional gain K iq  to the integrator  504 . The output from the integrator  504  is inputted to the adder  506 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     8 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     iq 
                   
                   = 
                   
                     
                       R 
                       a 
                     
                     
                       τ 
                       q 
                     
                   
                 
               
               
                 
                   ( 
                   8 
                   ) 
                 
               
             
           
         
       
     
     The adder  506  calculates the first q-axis voltage command value V q_dsh  by adding together the output of the gain block  502  and the output from the integrator  504 . As described above, the q-axis current control unit  406  can allow the transmission characteristic from the q-axis current command value i q2 * to the q-axis current i q  to match the normative response shown in the equation (9) below by setting each of the gain blocks  502  and  503  as shown in the equations (7) and (8) above. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     9 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     i 
                     q 
                   
                   = 
                   
                     
                       1 
                       
                         
                           
                             τ 
                             q 
                           
                           ⁢ 
                           s 
                         
                         + 
                         1 
                       
                     
                     · 
                     
                       i 
                       
                         q 
                         ⁢ 
                         2 
                       
                       * 
                     
                   
                 
               
               
                 
                   ( 
                   9 
                   ) 
                 
               
             
           
         
       
     
       FIG. 6  illustrates a diagram showing details of the d-axis current control unit  407  of the present embodiment. The d-axis current control unit  407  includes control blocks  601 ,  602 , and gain blocks  603 ,  604 , an integrator  605 , a subtractor  606 , and an adder  607 . 
     The control block  601  is a primary delay transmission characteristic (d-axis current transmission characteristic) 1/(τ d s+1), which simulates the response delay of the actual current (d-axis current i d ) for the d-axis current command value i d1 *. The control block  601  takes the d-axis current command value i d1 * as the input and outputs the d-axis current normative response i d_ref . The τd included in the transmission characteristic 1/(τ d s+1) is the time constant of the d-axis current normative response. 
     The control block  602  has the calculation function for the transmission characteristic s/(τ d s+1) that calculates the derivative of the d-axis current normative response i d_ref  for the d-axis current command value i d1 *. The control block  602  takes the d-axis current command value i d1 * as the input and outputs the d-axis current normative response derivative value, s·i d_ref . 
     The gain block  603  represents the proportional gain K pd , which is expressed by the following equation (10). The gain block  603  takes a deviation between the d-axis current command value i d1 * and the d-axis current i d  as the input, and outputs a value obtained by multiplying the inputted value by the proportional gain K pd  to the adder  607 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     10 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     
                       p 
                       ⁢ 
                       d 
                     
                   
                   = 
                   
                     
                       L 
                       d 
                       ′ 
                     
                     
                       τ 
                       d 
                     
                   
                 
               
               
                 
                   ( 
                   10 
                   ) 
                 
               
             
           
         
       
     
     The gain block  604  represents the proportional gain K id  which is expressed by the following equation (11). The gain block  604  takes a deviation between the d-axis current command value i d1 * and the d-axis current i d  as the input, and outputs a value obtained by multiplying the inputted value by the proportional gain K id  to the integrator  605 . The output of the integrator  605  is inputted to the adder  607 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     11 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     id 
                   
                   = 
                   
                     
                       R 
                       a 
                     
                     
                       τ 
                       d 
                     
                   
                 
               
               
                 
                   ( 
                   11 
                   ) 
                 
               
             
           
         
       
     
     the adder  607  calculates the first d-axis voltage command value v d_dsh  by adding together the output of the gain block  603  and the output of the integrator  605 . As described above, the d-axis current control unit  407  can allow the transmission characteristic from the d-axis current command values from i d1 * to the d-axis current i d  to match the normative response shown in the following equation (12) by setting each of the gain blocks  603  and  604  as shown in the above equations (10) and (11). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     12 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     i 
                     d 
                   
                   = 
                   
                     
                       1 
                       
                         
                           
                             τ 
                             d 
                           
                           ⁢ 
                           s 
                         
                         + 
                         1 
                       
                     
                     · 
                     
                       i 
                       
                         d 
                         ⁢ 
                         1 
                       
                       * 
                     
                   
                 
               
               
                 
                   ( 
                   12 
                   ) 
                 
               
             
           
         
       
     
       FIG. 7  shows the diagram illustrating the details of the f-axis current control unit  408  of the present embodiment. The f-axis current control unit  408  includes control blocks  701  and  702 , gain blocks  703  and  704 , an integrator  705 , a subtractor  706 , and an adder  707 . 
     The control block  701  has the calculation function for the transmission characteristic 1/(τ f s+1) with the primary delay that simulates the response delay of the actual current i f  for the f-axis current command value i f1 *. The control block  701  takes the f-axis current command value i f1 * as the input and outputs the f-axis current normative response i f_ref . Note that τ f  in the transmission characteristic 1/(τ f s+1) represents the f-axis current normative response time constant. 
     The control block  702  has the calculation function for the transmission characteristic s/(τ f s+1) that calculates the derivative of the f-axis current normative response i f_ref  for the f-axis current command value i f1 *. The control block  702  takes the f-axis current command value i f1 * as the input and outputs the f-axis current normative response derivative value s·i f_ref . 
     The gain block  703  represents the proportional gain K pf , which is expressed by the following equation (13). The gain block  703  takes a deviation between the f-axis current command value i f1 * and the f-axis current i f  as the input, and outputs a value obtained by multiplying the input value by the proportional gain K pf  to the adder  707 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     13 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     pf 
                   
                   = 
                   
                     
                       L 
                       f 
                       ′ 
                     
                     
                       τ 
                       f 
                     
                   
                 
               
               
                 
                   ( 
                   13 
                   ) 
                 
               
             
           
         
       
     
     The gain block  704  represents the integrator gain K if , and is expressed by the following equation (14). The gain block  704  takes a deviation between the f-axis current command value i f1 * and the f-axis current i f  as the input, and outputs a value obtained by multiplying the inputted value by the proportional gain K if  to the integrator  705 . The output of the integrator  705  is inputted to the adder  707 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     14 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     K 
                     if 
                   
                   = 
                   
                     
                       R 
                       f 
                     
                     
                       τ 
                       f 
                     
                   
                 
               
               
                 
                   ( 
                   14 
                   ) 
                 
               
             
           
         
       
     
     The adder  707  calculates the first f-axis voltage command value v f_dsh  by adding together the output of the gain block  703  and the output of the integrator  705 . As described above, by setting each of the gain blocks  703  and  704  as shown in the equations (13) and (14) above, the f-axis current control unit  408  can allow the transmission characteristic from the f-axis current command value i f1 * to the f-axis current i f  to match the normative response shown in the equation (15) below. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     15 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     i 
                     f 
                   
                   = 
                   
                     
                       1 
                       
                         
                           
                             τ 
                             f 
                           
                           ⁢ 
                           s 
                         
                         + 
                         1 
                       
                     
                     · 
                     
                       i 
                       
                         f 
                         ⁢ 
                         1 
                       
                       * 
                     
                   
                 
               
               
                 
                   ( 
                   15 
                   ) 
                 
               
             
           
         
       
     
     Referring back to  FIG. 4 , the explanation will be continued. The voltage command value calculation unit  409  compensates the first q-axis voltage command value v q_dsh , the first d-axis voltage command value v d_dsh , and the first f-axis voltage command value v f_dsh , which are the outputs of the q-axis current control unit  406 , the d-axis current control unit  407 , and the first f-axis voltage command value v f_dsh , by using the non-interference voltages v q_dcpl , v d_dcpl , and v f_dcpl , which are the outputs of the non-interference control unit  405  respectively (the compensations are made by adding in this embodiment). Then, the voltage command value calculation unit  409  outputs the second q-axis voltage command value v q * and the second d-axis voltage command value v d * obtained by the compensation to the coordinate conversion unit  410 , and outputs the second f-axis voltage command value v f * to the rotor PWM conversion unit  402 . 
     The coordinate conversion unit  410  performs a conversion from the orthogonal two-axes DC coordinate system (d-q axis) in the condition of rotating at an electric angular velocity ω re  to the three-phase AC coordinate system (uvw phase). Specifically, the coordinate conversion unit  410  calculates the voltage command values v u *, v v *, and v w * by performing the coordinate conversion processing using the following equation (16) from the inputted second d-axis voltage command value v d *, the second q-axis voltage command value v q *, and the electrical angle θ re ′ after look-ahead compensation. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     16 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     [ 
                     
                       
                         
                           
                             v 
                             u 
                             * 
                           
                         
                       
                       
                         
                           
                             v 
                             v 
                             * 
                           
                         
                       
                       
                         
                           
                             v 
                             w 
                             * 
                           
                         
                       
                     
                     ] 
                   
                   = 
                   
                     
                       
                         
                           
                             2 
                             3 
                           
                         
                         ⁡ 
                         
                           [ 
                           
                             
                               
                                 1 
                               
                               
                                 0 
                               
                             
                             
                               
                                 
                                   - 
                                   
                                     1 
                                     2 
                                   
                                 
                               
                               
                                 
                                   
                                     3 
                                   
                                   2 
                                 
                               
                             
                             
                               
                                 
                                   - 
                                   
                                     1 
                                     2 
                                   
                                 
                               
                               
                                 
                                   - 
                                   
                                     
                                       3 
                                     
                                     2 
                                   
                                 
                               
                             
                           
                           ] 
                         
                       
                       ⁡ 
                       
                         [ 
                         
                           
                             
                               
                                 cos 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   θ 
                                   re 
                                   ′ 
                                 
                               
                             
                             
                               
                                 
                                   - 
                                   sin 
                                 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   θ 
                                   re 
                                   ′ 
                                 
                               
                             
                           
                           
                             
                               
                                 sin 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   θ 
                                   re 
                                   ′ 
                                 
                               
                             
                             
                               
                                 cos 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   θ 
                                   re 
                                   ′ 
                                 
                               
                             
                           
                         
                         ] 
                       
                     
                     ⁡ 
                     
                       [ 
                       
                         
                           
                             
                               v 
                               d 
                               ⋆ 
                             
                           
                         
                         
                           
                             
                               v 
                               q 
                               * 
                             
                           
                         
                       
                       ] 
                     
                   
                 
               
               
                 
                   ( 
                   16 
                   ) 
                 
               
             
           
         
       
     
     Next, the details of the oscillation suppression process performed in step S 203  (see  FIG. 2 ) will be described. 
       FIG. 8  illustrates a control block diagram of an oscillation suppression calculation processing unit  2   b  provided in the controller  2  as a function unit. The controller  2  performs the oscillation control process pertaining to step S 203  using the oscillation suppression calculation processing unit  2   b.    
     The oscillation suppression calculation processing unit  2   b  includes a first current command value calculator  801 , a magnetic flux estimator  802 , a first torque command value calculator  803 , a second torque command value calculator  804 , and a second q-axis current command value calculator  805 . 
     The first current command value calculator  801  calculates the q-axis current command value i q1 *, the d-axis current command value i d1 *, and the f-axis current command value i f1 * from the input of the motor torque command value T m *, the motor rotation speed (machine angular velocity) ω rm , and the DC voltage V dc . The first current command value calculator  801  stores beforehand a map data which shows a relationship between each of the q-axis current command value i q2 *, the d-axis current command value i d1 *, and the f-axis current command value i f1 *, and the motor torque command value (basic torque command value) T m *, the motor rotation speed (machine angle speed) ω rm , and the DC voltage V dc , and calculates each value by referring to the map data. The calculated q-axis current command value i q2 * is outputted to the first torque command value calculator  803  and the d-axis current command value i d1 *, and the f-axis current command value i f1 * are outputted to the magnetic flux estimator  802 . 
       FIG. 9  illustrates a control block diagram of the magnetic flux estimator  802 . The magnetic flux estimator  802  includes a reluctance torque equivalent magnetic flux estimator  901 , a magnetic flux estimator  902 , and an adder  903 . 
     The reluctance torque equivalent magnetic flux estimator  901  calculates the reluctance torque equivalent magnetic flux estimate value φr{circumflex over ( )} using the d-axis current command value i d1 * as the input. The magnetic flux estimator  902  calculates the magnetic flux estimate value φf{circumflex over ( )} using the f-axis current command value i f1 * as the input. Then, the adder  903  calculates the magnetic flux estimate φ{circumflex over ( )} by adding together the reluctance torque equivalent magnetic flux estimate value φr{circumflex over ( )} and the magnetic flux estimate value φf{circumflex over ( )}. 
       FIG. 10  illustrates the control block diagram of the reluctance torque equivalent magnetic flux estimator  901 . The reluctance torque equivalent magnetic flux estimator  901  is composed of a phase-advancing compensator  1001  and a multiplier  1002 . 
     The phase-advancing compensator  1001  has a calculation function for a transmission characteristic (τ q s+1)/(τ d s+1), whereby the transmission characteristic of the primary delay (d-axis current transmission characteristic (see the control block  601 ) formed by simulating the d-axis current response delay is phase-advancing compensated by the q-axis current response. The phase-advancing compensator  1001  obtains the value by applying the phase-advancing compensation to the d-axis current command value i d1 * using the transmission characteristics (τ q s+1)/(τ d s+1), and outputs the obtained value to the multiplier  1002 . 
     The multiplier  1002  calculates the reluctance torque equivalent magnetic flux estimate φr{circumflex over ( )} by multiplying the output of the phase-advancing compensator  1001  by the difference L d −L q  between the d-axis inductance L d  and the q-axis inductance L q . For the d-axis inductance L d  and q-axis inductance L q , the values at any operating point (representative operating point) of the motor  4  may be used, or the values may be obtained with reference to the map data stored in advance. The reluctance torque generated in the rotor by the dq axis current i d  and i q  are represented by the following equation (17). Therefore, the item (L d −L q )i d  in the equation (17) below can be defined as the reluctance torque equivalent magnetic flux. Note that p n  is a pole number, e.g. the number of poles provided in the motor  4 .
 
[Equation 17]
 
 p   n ( L   d   −L   q ) i   d   i   q   (17)
 
     The oscillation suppression calculation processing unit  2   b  uses the reluctance torque equivalent magnetic flux estimate value φr{circumflex over ( )}, where the q-axis current response is phase advancing compensated by the phase-advancing compensator  1001 , such that the q-axis current response delay has been taken into consideration. With such calculation the q-axis current command value (second q-axis current command value) i q2 * can be calculated. 
       FIG. 11  shows the control block diagram of the magnetic flux estimator  902 . The magnetic flux estimator  902  includes a phase advancing compensator  1101  and a multiplier  1102 . 
     The phase advancing compensator  1101  has a calculation function for the transmission characteristic (τ q s+1)/(τ f s+1), where the q-axis current response is phase advancing compensated for the transmission characteristic of the primary delay (see the control block  701 ) formed by simulating the f-axis current response delay. The phase advancing compensator  1101  obtained by applying the phase advancing compensation using the transmission characteristics (τ q s+1)/(τ f s+1) to the f-axis current command value i f1 *, and outputs the obtained value to the multiplier  1102 . 
     The multiplier  1102  calculates the magnetic flux estimate value φf{circumflex over ( )} by multiplying the output of the phase advancing compensator  1101  by the mutual inductance M f  between the stator and the rotor. For the mutual inductance M f , a value at any operating point (representative operating point) of the motor  4  may be used, or it may be determined by referring to the map data stored in advance. 
     In the oscillation suppression calculation processing unit  2   b , the q-axis current command value i q2 *, with taking into consideration of the q-axis current response delay, can be calculated by using the magnetic flux estimate φf{circumflex over ( )} where the q-axis current response advancing compensation is performed by the phase advancing compensator  1101 . Hereinafter, the explanation will be continued referring back to  FIG. 8 . 
     The first torque command value calculator  803  calculates the first torque command value (torque command value before oscillation suppression) T m1 * by multiplying three of the q-axis current command value i q2 *, the magnetic flux estimate φ{circumflex over ( )}, and the poles number p n  of the motor  4 . The calculated first torque command value T m1 * is outputted to the second torque command value calculator  804 . 
     The second torque command value calculator  804  performs a so-called oscillation suppression control, which removes the natural oscillation frequency component of the vehicle&#39;s drive shaft torque transmission system, using the following equation (18) for the first torque command value T m1 *, thereby calculating the command value (final torque command value) T m2 *.
 
[Equation 18]
 
 T   m2   *   =G   INV ( s )· T   m1   (18)
 
     The derivation of the filter (transmission function) G inv (s) that removes the natural oscillation frequency component of the drive shaft torque transmission system of the vehicle will be described. First, the equation of motion of the vehicle will be described referring to  FIG. 12 . 
       FIG. 12  is the diagram in which the driving force transmission system of the vehicle is modeled to be a control block  601 , and each parameter in the figure is shown below. 
     J m : Motor inertia 
     J w : Drive wheel inertia (for 1 axis) 
     M: Vehicle mass 
     K d : Torsional rigidity of drive shaft (axel shaft) 
     K t : Coefficients for friction between tires and road surfaces 
     N at : Overall gear ratio 
     r: Tire load radius 
     ω m : Motor angular velocity 
     ω w : Drive wheel angle speed 
     T m : Motor torque 
     T d : Drive shaft torque 
     F: Driving force (2 axes) 
     V: Vehicle speed 
     The following equations of motion equations from (19) to (23) can be derived from  FIG. 12 . 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     19 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       J 
                       m 
                     
                     ⁢ 
                     
                       
                         ω 
                         . 
                       
                       m 
                     
                   
                   = 
                   
                     
                       T 
                       m 
                     
                     - 
                     
                       
                         T 
                         d 
                       
                       / 
                       
                         N 
                         
                           a 
                           ⁢ 
                           l 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   19 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     20 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     2 
                     ⁢ 
                     
                       J 
                       w 
                     
                     ⁢ 
                     
                       
                         ω 
                         . 
                       
                       w 
                     
                   
                   = 
                   
                     
                       T 
                       d 
                     
                     - 
                     rF 
                   
                 
               
               
                 
                   ( 
                   20 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     21 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     M 
                     ⁢ 
                     
                       V 
                       . 
                     
                   
                   = 
                   F 
                 
               
               
                 
                   ( 
                   21 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     22 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     T 
                     d 
                   
                   = 
                   
                     
                       K 
                       d 
                     
                     ⁢ 
                     
                       ∫ 
                       
                         
                           ( 
                           
                             
                               
                                 ω 
                                 m 
                               
                               
                                 N 
                                 
                                   a 
                                   ⁢ 
                                   l 
                                 
                               
                             
                             - 
                             
                               ω 
                               w 
                             
                           
                           ) 
                         
                         ⁢ 
                         dt 
                       
                     
                   
                 
               
               
                 
                   ( 
                   22 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     23 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   F 
                   = 
                   
                     
                       K 
                       t 
                     
                     · 
                     
                       ( 
                       
                         
                           r 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           
                             ω 
                             m 
                           
                         
                         - 
                         V 
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   23 
                   ) 
                 
               
             
           
         
       
     
     When the above equations from (19) to (23) are transformed with Laplace transformation and the transmission characteristic from the motor torque T m  to the motor angle speed ω m  are determined, they can be expressed by the following equations (24) and (25). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     24 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     ω 
                     m 
                   
                   = 
                   
                     
                       
                         G 
                         p 
                       
                       ⁡ 
                       
                         ( 
                         s 
                         ) 
                       
                     
                     · 
                     
                       T 
                       m 
                     
                   
                 
               
               
                 
                   ( 
                   24 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     25 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       G 
                       p 
                     
                     ⁡ 
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       1 
                       s 
                     
                     · 
                     
                       
                         
                           
                             b 
                             3 
                           
                           ⁢ 
                           
                             s 
                             3 
                           
                         
                         + 
                         
                           
                             b 
                             2 
                           
                           ⁢ 
                           
                             s 
                             2 
                           
                         
                         + 
                         
                           
                             b 
                             1 
                           
                           ⁢ 
                           s 
                         
                         + 
                         
                           b 
                           0 
                         
                       
                       
                         
                           
                             a 
                             3 
                           
                           ⁢ 
                           
                             s 
                             3 
                           
                         
                         + 
                         
                           
                             a 
                             2 
                           
                           ⁢ 
                           
                             s 
                             2 
                           
                         
                         + 
                         
                           
                             a 
                             1 
                           
                           ⁢ 
                           s 
                         
                         + 
                         
                           a 
                           0 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   25 
                   ) 
                 
               
             
           
         
       
     
     Where a 3 , a 2 , a 1 , a 0 , b 3 , b 2 , b 1 , and b 0  in the equations (24) and (25) are represented by the following equation (26), respectively.
 
[Equation 26]
 
α 3 =2 J   m   J   w   M  
 
α 2   =K   t   J   m (2 J   w   +r   2   M )
 
α 1   =K   d   M ( J   m +2 J   w   /N   2 )
 
α 0   =K   d   K   t ( J   m +2 J   w   /N   2   +r   2   M/N   2 )
 
 b   3 =2 J   w   M  
 
 b   2   =K   t (2 J   w   +r   2   M )
 
 b   1   =K   d   M  
 
 b   0   =K   d   K   t   (26)
 
     When the equation (25) is arranged and simplified, G p  (s) can be expressed as the following equation (27), where, ζ p  and ω p  in the equation (27) are a suppression coefficient and an inherent oscillation frequency of the drive shaft torsional oscillation system, respectively. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     27 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       G 
                       p 
                     
                     ⁡ 
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       1 
                       s 
                     
                     · 
                     
                       
                         
                           ( 
                           
                             s 
                             + 
                             β 
                           
                           ) 
                         
                         · 
                         
                           ( 
                           
                             
                               
                                 b 
                                 2 
                                 ′ 
                               
                               ⁢ 
                               
                                 s 
                                 2 
                               
                             
                             + 
                             
                               
                                 b 
                                 1 
                                 ′ 
                               
                               ⁢ 
                               s 
                             
                             + 
                             
                               b 
                               0 
                               ′ 
                             
                           
                           ) 
                         
                       
                       
                         
                           ( 
                           
                             s 
                             + 
                             α 
                           
                           ) 
                         
                         · 
                         
                           ( 
                           
                             
                               s 
                               2 
                             
                             + 
                             
                               2 
                               ⁢ 
                               
                                 ζ 
                                 p 
                               
                               ⁢ 
                               
                                 ω 
                                 p 
                               
                               ⁢ 
                               s 
                             
                             + 
                             
                               ω 
                               p 
                               2 
                             
                           
                           ) 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   27 
                   ) 
                 
               
             
           
         
       
     
     If the ideal model G m (s) indicating the response target of the motor rotation speed to the inputted torque is set to the vehicle in the following equation (28), the transmission function G inv  (s) can be represented by the following equation (29), where, ζ m  and ω m  in the equations (28), (29) are the suppression coefficient and the inherent oscillation frequency of the drive shaft torsional oscillation system, respectively. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     28 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       G 
                       m 
                     
                     ⁡ 
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       1 
                       s 
                     
                     · 
                     
                       
                         
                           ( 
                           
                             s 
                             + 
                             β 
                           
                           ) 
                         
                         · 
                         
                           ( 
                           
                             
                               
                                 b 
                                 2 
                                 ′ 
                               
                               ⁢ 
                               
                                 s 
                                 2 
                               
                             
                             + 
                             
                               
                                 b 
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                               ⁢ 
                               s 
                             
                             + 
                             
                               b 
                               0 
                               ′ 
                             
                           
                           ) 
                         
                       
                       
                         
                           ( 
                           
                             s 
                             + 
                             α 
                           
                           ) 
                         
                         · 
                         
                           ( 
                           
                             
                               s 
                               2 
                             
                             + 
                             
                               2 
                               ⁢ 
                               
                                 ζ 
                                 m 
                               
                               ⁢ 
                               
                                 ω 
                                 m 
                               
                               ⁢ 
                               s 
                             
                             + 
                             
                               ω 
                               m 
                               2 
                             
                           
                           ) 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   28 
                   ) 
                 
               
             
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     29 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       G 
                       INV 
                     
                     ⁡ 
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       
                         
                           G 
                           m 
                         
                         ⁡ 
                         
                           ( 
                           s 
                           ) 
                         
                       
                       
                         
                           G 
                           p 
                         
                         ⁡ 
                         
                           ( 
                           s 
                           ) 
                         
                       
                     
                     = 
                     
                       
                         
                           s 
                           2 
                         
                         + 
                         
                           2 
                           ⁢ 
                           
                             ζ 
                             p 
                           
                           ⁢ 
                           
                             ω 
                             p 
                           
                           ⁢ 
                           s 
                         
                         + 
                         
                           ω 
                           p 
                           2 
                         
                       
                       
                         
                           s 
                           2 
                         
                         + 
                         
                           2 
                           ⁢ 
                           
                             ζ 
                             m 
                           
                           ⁢ 
                           
                             ω 
                             m 
                           
                           ⁢ 
                           s 
                         
                         + 
                         
                           ω 
                           m 
                           2 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   29 
                   ) 
                 
               
             
           
         
       
     
     In addition, the known method disclosed in JP5900609B can be also applied to the oscillation suppression process of this embodiment as the method of removing the natural oscillation frequency component of the drive shaft torque transmission system of the vehicle by adjusting the natural oscillation frequency component in consideration of the influence of a backlash of gears in situations where the vehicle accelerates from a coast stop or deceleration, 
     In addition, the second q-axis current command value calculator  805  shown in  FIG. 8  takes the second torque command value T m2 * outputted from the second torque command value calculator  804  and the magnetic flux estimate φ{circumflex over ( )} outputted from the magnetic flux estimator  802  as the inputs. Later, the second q-axis current command value calculator  805  calculates the q-axis current command value (second q-axis current command value) i q2 * by using the following equation (30). The calculated q-axis current command value i q2 * is inputted to the q-axis current control unit  406  of the current control calculation processing unit  2   a  shown in  FIG. 4 . As described above, the magnetic flux estimate φ{circumflex over ( )} is calculated based on the d-axis current command value i d1 * set according to the torque command value T m * based on the vehicle information and the f-axis current command value i f1 *. In other words, the q-axis current command value i q2 * of the present embodiment is calculated by compensating the q-axis current command value i q2 * set for the torque command value T m *, taking into account the d-axis current command value i d1 * and the f-axis current command value i f1 *. As the result, the oscillation suppression calculation processing unit  2   b  can suppress the torsional oscillation of the drive shaft torque transmission system in consideration of the influence of the reluctance torque generated for the d-axis current command value i d1 * and the magnetic flux generated for the f-axis current command value i n *.
 
[Equation 30]
 
 i   q2   *=T   m2 */( p   n ·{circumflex over (ϕ)})  (30)
 
     In the following statements referring to  FIG. 13 , the effect by the control method (oscillation suppression processing) of the electric vehicle of the first embodiment described above will be explained. 
       FIG. 13  is the time chart showing the control results of the present embodiment. The horizontal axis represents time, and the vertical axis represents the motor torque command value [Nm], the vehicle&#39;s forward and backward acceleration [m/s2], and the f-axis voltage [V] in the order from top left, and the q-axis current command value [A], the d-axis current command value [A], and the f-axis current command value [A] in the order from top right. The solid lines in the figure show the present embodiment, and the dotted lines show the control by the conventional technique (conventional example). 
       FIG. 13  represents a scene when the motor torque command value is changed in a step shape accelerated (rising) at the time t 1  from the state in which the vehicle stopped. 
     In the control of the present embodiment in this time chart, the f-axis current normative response time constant (see the control block  701  (τ f )) is set to the value where f-axis voltage saturation does not occur. When this embodiment is applied, since the motor torque that suppresses drive shaft torsional oscillation is realized (see  FIG. 8 ) by the q-axis current command value i q2 * calculated taking into account the d-axis current i d  and the f-axis current, the acceleration oscillation in front and the rear of the vehicle is suppressed as shown by the solid lines in the figure. 
     On the other hand, in the conventional control method, since the f-axis current is not considered when the q-axis current command value is calculated, the drive shaft torsional oscillation is generated, and the acceleration oscillation in the direction of front/rear of the vehicle is generated as shown by the dotted lines. 
     As described above, the control method of the electric vehicle of the first embodiment is the control method for controlling the currents of the electric vehicle driven by the field magnet winding type synchronous motor  4  having the stator winding through which the stator current flows and the rotor winding through which the rotor current flows. In this control method, the basic torque command value T m * is set based on the vehicle information. The d-axis current command value i d1 *, the first q-axis current command value i q2 * for the stator current, and the f-axis current command value i f1 * for the rotor current are calculated based on the basic command value and the vehicle information. The estimated flux value φ{circumflex over ( )}, which is the estimated value of magnetic flux generated in the rotor, is calculated based on the d-axis current command value i d1 * and the f-axis current command value i f1 *. Then, the final torque command value T m2 * is calculated based on the first q-axis current command value i q2 * and the estimated flux value φ{circumflex over ( )}, and the second q-axis current command value i q2 * is calculated based on the magnetic flux estimate φ{circumflex over ( )} and the final torque command value T m2 *. The stator current and the rotor current are controlled based on the second q-axis current command value i q2 *, the d-axis current command value i d1 *, and the f-axis current command value i f1 *. This allows the second q-axis current command value i q2 * to be calculated in consideration of the d-axis current command value i d1 * and the f-axis current command value i f1 *, such that it is possible to apply oscillation suppression control to suppress torsional oscillation in the drive shaft torque transmission system of the electric vehicle using the field magnet winding type synchronous motor  4  as the drive source, taking into consideration of the effect of the reluctance torque generated by the d-axis current command value i d1 * and the magnetic flux generated by the f-axis current command value i f1 *. 
     Further, according to the control method of the electric vehicle of the first embodiment, the final torque command value T m2 * can be calculated by applying oscillation suppression control to suppress torsional oscillation of the drive shaft torque transmission system using the filter G inv (s) that removes the natural oscillation frequency components of the drive shaft torque transmission system of the electric vehicle for the torque command value T m1 * before the oscillation suppression, which is calculated based on the first q-axis current command value i q2 * and the magnetic flux estimate value φ{circumflex over ( )}. This allows applying oscillation suppression control that suppresses the torsional oscillation of the drive shaft torque transmission system of the electric vehicle, so that it is possible to suppress the generation of torsional oscillation of the drive shaft torque transmission system of the electric vehicle using the field magnet winding type synchronous motor  4  as the driving source. 
     In addition, according to the control method of the electric vehicle of the first embodiment, the second q-axis current command value i q2 * is calculated by dividing the final torque command value T m2 * by the magnetic flux estimate φ{circumflex over ( )}. Thereby, the q-axis current command value i q2 * that realizes the final torque command value T m2 * applied to oscillation control can be calculated. 
     Furthermore, according to the control method of the electric vehicle of the first embodiment, the field magnetic flux estimate value φf{circumflex over ( )}, which is the estimated value of the magnetic flux of the rotor, is calculated based on the f-axis current command value i f1 *, the equivalent magnetic flux estimate value φr{circumflex over ( )} of the reluctance torque generated in the rotor based on the d-axis current command value i d1 * is calculated, and the magnetic flux estimate φ{circumflex over ( )} is calculated by adding together the magnetic flux estimate φf{circumflex over ( )} and the equivalent magnetic flux estimate value φr{circumflex over ( )}. Thereby, considering the influence of the d-axis current i d  and the f-axis current i f , the q-axis current command value i q2 * that realizes the final torque command value T m2 * applied to the oscillation suppression control can be calculated. 
     In addition, according to the control method of the electric vehicle of the first embodiment, the magnetic flux estimate φ{circumflex over ( )} is calculated using the configured transmission characteristics (the control block  1101 ) to phase-advancing compensate the q-axis current response for the f-axis current command value i f1 * of the f-axis current i f  that constitutes the rotor current for. Thereby, the q-axis current command value i q2 *, where the q-axis current response delay for the d-axis current i d  is taken into consideration can be calculated. 
     Furthermore, the control method of the electric vehicle of the first embodiment has the function of the f-axis current transmission characteristic for calculating the transmission&#39;s primary delay. Thereby, the f-axis current response which does not cause an f-axis voltage saturation can be appropriately simulated. 
     In addition, according to the control method of the electric vehicle of the first embodiment, the equivalent magnetic flux estimate φr{circumflex over ( )} can be calculated by using the transmission characteristic constituted for performing phase-advancing compensation of the q-axis current response for the d-axis current transmission characteristic that simulates the response delay for the d-axis current command value of the d-axis current i d  that constitutes the rotor current. Thereby the q-axis current command value in consideration of the q-axis current delay for the f-axis current i f  can be calculated. 
     Second Embodiment 
     The second embodiment of the control method for the electric vehicle will be described. In the first embodiment, the method was explained that when the non-interference control by the non-interference control unit  405  functions ideally, the d-axis, q-axis, and f-axis voltage-to-current characteristics show the primary delay as shown in Equations (4), (5), and (6) above, respectively. However, when the f-axis voltage has been saturated, the f-axis current response does not match the normative response of the primary delay. The control method of the electric vehicle of this embodiment is the control method applied on the presumption that the f-axis current i f  is controlled in consideration of the f-axis voltage saturation, and in particular, the configuration of a magnetic flux estimator  802  provided by the oscillation suppression calculation processing unit  2   b  differs from the first embodiment. 
     Prior to the description of the magnetic flux estimator  802  of this embodiment, the method of controlling the f-axis current i f  will be described in consideration of the f-axis voltage saturation. Since the controls on the d-axis and q-axis are the same as that on the f-axis, the explanations for the d-axis and q-axis are omitted, and only the control on the f-axis will be explained below. 
       FIG. 14  illustrates the example of the configuration of the motor control system  200  of the second embodiment. The motor control system  200  of this embodiment differs from the first embodiment in that the power supply voltage V dc  of the battery  1  and the non-interference voltage V f_dcpl , which is the output of the non-interference control unit  405 , are inputted to the f-axis current control unit  408 . 
     The details of the f-axis current control unit  408  will be described using  FIG. 15 .  FIG. 15  illustrates a control block diagram of the f-axis current control unit  408 . 
     In the f-axis current control unit  408 , the first f-axis voltage command value v f_dsh  is calculated so that the f-axis current i f  inputted from the A/D converter  411  follows the f-axis current command value i f  with a desired responsiveness without steady deviation. Further, the f-axis current control unit  408  calculates the f-axis current normative response i f_ref  and the f-axis current normative response derivative value s·i f_ref , which are used in later processing. The f-axis current control unit  408  includes an f-axis F/F (feed-forward) compensator  201 , an f-axis F/B compensator  202 , an f-axis robust compensator  203 , and an f-axis limit processing unit  204 , each of which will be described in detail below. 
     The f-axis F/F compensator  201  takes the f-axis current command value i f*  as the input and calculates, in addition to the f-axis F/F compensation voltage v f_ff , the f-axis current normative response i f_ref  and the f-axis current normative response derivative value s·i f_ref . The f-axis F/F compensator  201  outputs the f-axis current normative response i f_ref  and the f-axis current normative response derivative value s i f_ref  to the non-interference control unit  405 , and also outputs the f-axis current normative response i f_ref  to the f-axis F/B compensator  202 . The details of the f-axis F/F compensator  201  will be described later using  FIG. 16 . Although not shown in the figure, the power supply voltage V dc  outputted from the battery  1  and the non-interference voltage V f_dcpl  are inputted to the f-axis F/F compensator  201 . 
     The f-axis F/B compensator  202  is a compensator that performs general feedback compensation. The f-axis F/B compensator  202  performs F/B processing to negatively feedback the f-axis current when measured by the current sensor  7  to the f-axis current normative response if ref calculated in the f-axis F/F compensator  201 , and calculates the f-axis F/B compensation voltage v f_fb . The f-axis F/B compensator  202  outputs the f-axis F/B compensation voltage v f_fb  to the adder  205 . The details of the f-axis F/B compensator  202  will be described later using  FIG. 21 . The f-axis F/B compensator  202  is an example of the block that performs the F/B compensation step. 
     The f-axis robust compensator  203  calculates, based on the first f-axis voltage command value v f_dsh  calculated in the f-axis limit processing unit  204  described below and finally outputted from the f-axis current control unit  408  and the f-axis current i f , the f-axis robust compensation voltage v f_rbst  to ensure the robustness of the system. The f-axis robust compensator  203  outputs the f-axis robust compensation voltage v f_rbst  to the adder  206 . The details of the f-axis robust compensator  203  will be described later using  FIG. 22 . 
     The two adders  205  and  206  are provided in the former stage of the f-axis limit processing unit  204 . The f-axis F/B compensation voltage v f_fb  is added to the f-axis F/F compensation voltage v f_ff  calculated in the f-axis F/F compensator  201  by the adder  205 , and further, the f-axis robust compensation voltage v f_rbst  is added by the adder  206 . The final added value is then inputted to the f-axis limit processing unit  204 . Accordingly, the f-axis limit processing unit  204  takes the values calculated by adding the sum of the f-axis F/B compensation voltage v f_fb , which is the F/B compensation value, and the f-axis robust compensation voltage v f_rbst , which is the F/B robust compensation value to the f-axis F/F compensation voltage v f_ff , which is the F/F command value. 
     Then, the f-axis limit processing unit  204  limits the input voltage command value and calculates the first f-axis voltage command value v f_dsh . The f-axis limit processing unit  204  outputs the f-axis voltage command value v f_dsh  to the voltage command value calculation unit  409  and the f-axis robust compensator  203 . In the f-axis limit processing unit  204 , the same processing is performed as in an f-axis limit processing unit  303  described below using  FIGS. 19 and 20 . 
     Next, the detailed configuration of the f-axis F/F compensator  201  will be described using  FIG. 16 .  FIG. 16  illustrates a detailed block diagram of the f-axis F/F compensator  201 . The f-axis F/F compensator  201  has an f-axis current model  301 , an f-axis current pseudo-F/B model  302 , and the f-axis limit processing unit  303 . 
     The f-axis current model  301  is a filter that models the normative response characteristics from the f-axis voltage to the f-axis current. The f-axis current model  301  calculates the f-axis current normative response i f_ref , which is the normative response, by filtering processing using the normative response model from the voltage to the current in the f-axis for f-axis compensation voltage v f_ff  outputted from the f-axis limit processing unit  303  described later, and outputs the response to the f-axis F/B compensator  202 . The f-axis current model  301  also outputs the f-axis current normative response derivative value s i f_ref , which is the derivative of the f-axis current normative response i f_ref , to the non-interference control unit  405  for use in later processing. The details of the f-axis current model  301  will be described later using  FIG. 17 . 
     In the f-axis current pseudo-F/B model  302 , the f-axis current normative response i i_ref  outputted from the f-axis current model  301  is negatively fed back to the f-axis current command value i f * calculated by the current command value calculator  113 . The f-axis current pseudo-F/B model  302  calculates the pseudo-FB voltage command value v f_pse_fb  in order to make the f-axis current normative response i i_ref  follow the f-axis current command value i f * with the desired responsiveness without steady deviation, and outputs the value to the f-axis limit processing unit  303 . The details of the f-axis current pseudo-F/B model  302  will be described later using  FIG. 18 . 
     The f-axis limit processing unit  303  limits the pseudo-FB voltage command value v f_pse_fb  outputted from the f-axis current pseudo-F/B model  302 , calculates the f-axis F/F compensation voltage v f_ff , and outputs the value to the adder  205  and the f-axis current model  301 . The details of the f-axis limit processing unit  303  will be described later using  FIGS. 19 and 20 . 
     Although not shown in the figures, the power supply voltage V dc  outputted from the battery  1  and the non-interference voltage V f_dcpl  outputted from the non-interference control unit  405  are inputted to the f-axis limit processing unit  303 . As shown in  FIG. 15 , the f-axis F/F compensation voltage v f_ff  outputted from the f-axis limit processing unit  303  passes through the adder  205 , the adder  206 , and the f-axis limit processing unit  204  to calculate the first f-axis voltage command value v f_dsh . In other words, the combination of the adder  205 , the adder  206 , and the f-axis limit processing unit  204  are examples of the sets constituting the block configuration that executes the first f-axis voltage command value calculation step. 
     Therefore, in the f-axis F/F compensator  201 , a F/B system in which the measured f-axis current i f  is negatively fed back is not provided, but a pseudo-F/B model in which the f-axis current normative response i f_ref  calculated in the f-axis current model  301  is negatively fed back for the f-axis current pseudo-F/B model  302  is provided. By realizing the pseudo-F/B system in this way, a F/B control with poor response can be avoided, thus the responsiveness is improved. 
     Furthermore, as shown in  FIG. 14 , since the f-axis voltage v f  is generated by the battery  1 , the upper limit of the f-axis voltage v f  is limited and saturated by the supply voltage V dc  of the battery  1 . Therefore, the f-axis limit processing unit  303 , which models saturation at the power supply voltage V dc , is provided to limit the first f-axis voltage command value v f_dsh  and calculate the f-axis F/F compensation voltage v f_ff . The f-axis F/F compensation voltage v f_ff  in which voltage saturation is taken into account is returned to the f-axis current pseudo-F/B model  302 , thereby improving the accuracy of the rotation control. 
     Next, a detailed configuration of the f-axis current model  301  will be described using  FIG. 17 .  FIG. 17  is a detailed block diagram of the f-axis current model  301 . The f-axis current model  301  has a multiplier  1401 , a subtractor  1402 , a divider  1403 , and an integrator  1404 . 
     The multiplier  1401  multiplies the rotor winding resistance R f  by the f-axis current normative response i f_ref , which is one of the final outputs of the f-axis current model  301  and is output from the integrator  1404  described below, and outputs the result of the multiplication to the subtractor  1402 . The result of this multiplication corresponds to the voltage value of the normative response. 
     The subtractor  1402  subtracts the voltage value of the normative response output from the multiplier  1401  from the f-axis F/F compensation voltage v f_ff  output from the f-axis limit processing unit  303 , and outputs the subtracted value to the divider  1403 . 
     The divider  1403  divides by a f-axis dynamic inductance L f ′ for the subtracted value calculated by the subtractor  1402 , and outputs the result of the division to the non-interference control unit  405  and to the integrator  1404 . In this way, the f-axis current normative response derivative value s·i f_ref  is calculated. 
     The integrator  1404  calculates the f-axis current normative response i f  ref by integrating the f-axis current normative response derivative value s i f_ref  output from the divider  1403 , and outputs the f-axis current normative response i f_ref  to the non-interference control unit  405 , the f-axis F/B compensator  202 , and the multiplier  1401 . 
     Therefore, in the f-axis current model  301 , the f-axis current normative response i f_ref , which is one of the final outputs, is multiplied by the rotor winding resistance R f  by the multiplier  1401  and the multiplication result is negatively fed back to the f-axis F/F compensation voltage v f_ff , which is used as the input. By dividing the resultant value of this negative feedback by the f-axis dynamic inductance L f ′ with the divider  1403 , the f-axis current normative response i f_ref  based on the f-axis F/F compensation voltage v f_ff  and its derivative value s·i f_ref  can be obtained. 
     Next, a detailed configuration of the f-axis current pseudo-F/B model  302  will be described using  FIG. 18 .  FIG. 18  illustrates a detailed block diagram of the f-axis current pseudo-F/B model  302 . The f-axis current pseudo-F/B model  302  has a filter  1501 , a filter  1502 , and a subtractor  1503 . 
     The filter  1501  has the function of multiplying the f-axis current command value i f * output from the current command value calculator  113  by a gain G af , and outputs the filtered value to the subtractor  1503 . 
     The filter  1502  has the function of multiplying the f-axis current normative response i f_ref  output from the f-axis current model  301  by a gain G bf , and outputs the filtered value to the subtractor  1503 . 
     The subtractor  1503  calculates the pseudo-F/B voltage command value v f_pse_fb  by subtracting the output value of the filter  1502  from the output value of the filter  1501 , and outputs the pseudo-FB voltage command value v f_pse_fb  to the f-axis limit processing unit  303 . In other words, the pseudo-F/B control is configured by the negative feedback of the f-axis current normative response i f_ref , which is not a measured value. 
     However, the gain G af  and the gain G bf  can be shown as the following Equation (31). The parameter τ f  represents the f-axis current control norm response time constant (f-axis current norm response time constant). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     31 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       G 
                       
                         a 
                         ⁢ 
                         f 
                       
                     
                     = 
                     
                       
                         L 
                         f 
                         ′ 
                       
                       
                         τ 
                         f 
                       
                     
                   
                   , 
                   
                     
                       G 
                       
                         b 
                         ⁢ 
                         f 
                       
                     
                     = 
                     
                       
                         
                           L 
                           f 
                           ′ 
                         
                         - 
                         
                           
                             τ 
                             f 
                           
                           ⁢ 
                           
                             R 
                             f 
                           
                         
                       
                       
                         τ 
                         f 
                       
                     
                   
                 
               
               
                 
                   ( 
                   31 
                   ) 
                 
               
             
           
         
       
     
     With this configuration described above, in the f-axis current pseudo-F/B model  302 , pseudo-F/B control can be realized for the f-axis current command value i f*  by using the f-axis current normative response i f_ref  as the F/B component instead of the actually measured f-axis current if. 
     Next, the detailed configuration of the f-axis limit processing unit  303  will be described using  FIG. 19 .  FIG. 19  illustrates a detailed block diagram of the f-axis limit processing unit  303 . The f-axis limit processing unit  303  has a comparator  1601 , an inverter  1602 , a comparator  1603 , and subtractors  1604  and  1605 . 
     In the subtractor  1604 , which is provided in the former stage of the comparator  1601 , a subtraction value is obtained by subtracting the f-axis non-interference voltage v f_dc  outputted from the non-interference control unit  405  from the power supply voltage V dc  of the battery  1 . Then, the comparator  1601  compares the pseudo-FB voltage command value V f_pse_fb , which is the output value from the f-axis current pseudo-F/B model  302 , with the subtraction value in the subtractor  1604 , and outputs a smaller value to the comparator  1603 . 
     The inverter  1602  inverts the sign of the supply voltage V dc . 
     The subtractor  1605  is provided in the former stage of the comparator  1603 . The subtractor  1605  obtains the subtraction value by subtracting the f-axis non-interference voltage V f_dcpl  outputted from the non-interference control unit  405  from the output of the inverter  1602 . Then, the comparator  1603  compares the output value of the comparator  1601  with the subtracted value in the subtractor  1605 , and outputs the larger value thereof to f-axis current model  301  and to the adder  205 . 
     With this configuration, in the f-axis limit processing unit  303 , in order to obtain enough margin to add the f-axis non-interference voltage v f_dcpl  to the pseudo-FB voltage command value V f_pse_fb , which is the output value of the f-axis current pseudo-F/B model  302 , a limiting process is performed. In the limiting process, limitation based on the supply voltage V dc  in which the f-axis non-interference voltage V f_dcpl  is negatively offset, is performed. Specifically, the upper limit is “V dc_ V f_dcpl   ”  and the lower limit is “−V dc −V f_dcpl ” in the limiting process. 
     The f-axis limit processing unit  303  may be configured as shown in  FIG. 20 .  FIG. 20  illustrates another example of the detailed block diagram of the f-axis limit processing unit  303 . In this example, the f-axis limit processing unit  303  has a comparator  1701 , an inversion unit  1702 , a comparator  1703 , a subtractor  1704 , and an adder  1705 . 
     The adder  1705  is provided in the former stage of the comparator  1701 , and the adder  1705 , and adds the f-axis non-interference voltage V f_dcpl  outputted from the non-interference control unit  405  to the f-axis the pseudo-FB voltage command value v f_pse_fb  output from the f-axis current pseudo-F/B model  302 . Then, comparator  1701  compares the supply voltage V dc  of the battery  1  with the result of the addition in the adder  1705 , and outputs a smaller value thereof to the comparator  1703 . 
     The inversion unit  1702  inverses the sign of the supply voltage V dc . 
     The comparator  1703  compares the output from the comparator  1701  with the output from the inversion unit  1702  and outputs a larger value to the subtractor  1704 . 
     The subtractor  1704  subtracts the f-axis non-interference voltage v f_dcpl  outputted from the non-interference control unit  405  from the output value of the comparator  1703  to calculate the f-axis F/F compensation voltage v f_ff . The subtractor  1704  outputs the f-axis F/F compensation voltage v f_ff  to the f-axis current model  301  and the adder  205  constituting the f-axis current control unit  408 . 
     Even with this configuration, in the f-axis limit processing unit  303 , in order to obtain enough margin to add the f-axis non-interference voltage v f_dcpl  to the pseudo-FB voltage command value v f_pse_fb , which is the output value of the f-axis current pseudo-F/B model  302 , a limiting process is performed. In the limiting process, limitation based on the supply voltage V de  in which the f-axis non-interference voltage V f_dcpl  is negatively offset, is performed. Specifically, the upper limit is “V dc_ V f_dcpl   ”  and the lower limit is “−V dc -V f_dcpl ” in the limiting process. 
     Next, details of the f-axis F/B compensator  202  will be described.  FIG. 21  illustrates a detailed block diagram of the f-axis F/B compensator  202 . The f-axis F/B compensator  202  has a block  1801 , a multiplier  1802 , and a subtractor  1803 . 
     The block  1801  represents a delay filter, which delays the control system by a certain amount of latent time. The block  1801  delays the f-axis current normative response i f_ref  for an input of a f-axis current normative response i f  output from the f-axis F/F compensator  201 , calculates a f-axis current normative response after wasted time processing i _ref ′ in order to match the phase of the f-axis current normative response i _ref  and the f-axis current i f , and outputs the normative response to the subtractor  1803  provided in the first stage of the multiplier  1802 . The latent time L of the control system corresponds to the control operation delay. The block  1801  is an example of the block that performs the delay step. 
     The subtractor  1803  calculates the subtraction result by subtracting the f-axis current i f_ref ′ output from the A/D converter  107  from the f-axis current normative response after wasted time processing i _ref ′ output from the block  1801 . 
     The multiplier  1802  calculates the f-axis F/B compensation voltage v f_fb  by multiplying a f-axis F/B gain K f  with the subtraction result in the subtractor  1803  as an input, and outputs the f-axis F/B compensation voltage v f_fb  to the adder  205 . The value of the f-axis F/B gain K f  is determined by an experimental adjust so that the stability of the gain margin, phase margin, etc. satisfies a predetermined standard. 
     With this configuration, the f-axis F/B compensation voltage v f_fb  based on the f-axis current i f  is calculated in the f-axis F/B compensator  202 . 
       FIG. 22  illustrates a detailed block diagram of the f-axis robust compensator  203 . The f-axis robust compensator  203  includes a block  1901 , a block  1902 , a block  1903 , and a subtractor  1904 . 
     The block  1901  performs filter processing on the f-axis current i f  outputted from the A/D converter  107  as the input to calculate the first f-axis voltage estimate v f_est1 , and outputs the estimate to the subtractor  1904 . The block  1901  is a delay filter with the characteristics of (L f ′·s+R f )/(τ h_f ·s+1), including the low-pass filter 1/(τ h_f ·s+1) of the block  1903  described below. 
     The block  1902  represents the same delay filter as the block  1801 . The block  1902  calculates a second f-axis voltage estimate v f_est2  by delaying the first f-axis voltage command value v f_dsh  outputted from the f-axis limit processing unit  204  by the latent time L that the control system has. Then, the block  1902  outputs the second f-axis voltage estimate v f_est2  to the block  1903 . 
     The block  1903  represents a low pass filter with a characteristic of 1/(τ h_f ·s+1). The block  1903  performs low-pass filter processing on the second f-axis voltage estimate v f_est2  outputted from the block  1902  to calculate a third f-axis voltage estimate v f_est3 . Then, the block  1903  outputs the third f-axis voltage estimate v f_est3  to the subtractor  1904 . 
     The subtractor  1904  subtracts the first f-axis voltage estimate v f_est1  from the third f-axis voltage estimate v f_est3  to calculate the f-axis robust compensation voltage v f_rbst  and send the voltage to the adder  206 . 
     In this manner, the first f-axis voltage command value v f_dsh  is subjected to the processing of the block  1901 , which is the delay filter, and the block  1903 , which is a low-pass filter, and subtracting the first f-axis voltage estimate v f_est1  based on the measured value, such that the f-axis robust compensation voltage v f_rbst  is calculated in order to improve stability further. 
       FIG. 23  is a flowchart showing the control processing of the motor  4  explained by using the above  FIGS. 14 to 22 . These controls are performed by the controller  2  by executing a predetermined program. 
     In step S 1 , the current values (u-phase current i us , v-phase current i vs , and f-axis current i f ), and the electric angle θ re  of the motor  4  are obtained by the A/D converter  411 . 
     In step S 2 , based on the electric angle obtained in step S 1 , the mechanical angular velocity ω rm  as the motor rotation speed, and the electric angular velocity ω rθ  as the motor rotation speed, are calculated based on the electric angle θ re  obtained in step S 1 . 
     In step S 3 , the look-ahead compensation unit  403  calculates the after forward-read compensation electric angle θ re ′, based on the electric angle calculated in step S 2 . 
     In step S 4 , the coordinate transformation unit  404  calculates the d-axis current i d  and the q-axis current i q  based on the u-phase current i u  and the v-phase current i v  calculated in step S 1 . 
     In step S 5 , the d-axis current command value i d *, the q-axis current command value i q *, and the f-axis current command value i f * are calculated based on the motor speed ω rm , the torque command value T*, and the power supply voltage V dc . 
     In step S 6 , a q-axis current control unit  406 , a d-axis current control unit  407 , the f-axis current control unit  408  calculate the first d-axis voltage command value v q_dsh , the d-axis current normative response i d_ref , the d-axis current normative response derivative value s·i d_ref , the first q-axis voltage command value v q_dsh , the q-axis current normative response ref, the first f-axis voltage command value v f_dsh , and the f-axis current normative response i _ref , and the f-axis current normative response derivative value s·i f_ref  are calculated. 
     In step S 7 , the non-interference control unit  405  calculates the electric angular velocity ω re  calculated in step S 2 , and the d-axis current normative response i d_ref , the derivative of the d-axis current normative response s·i d_ref , q-axis current normative response i q_ref , the f-axis current normative response if ref, and the f-axis current normative response derivative value s i f_ref  calculated in step S 6 , the non-interference voltages v d_dcpl , v q_dcpl , and v f_dcpl . 
     In step S 8 , the voltage command value calculation unit  409  calculates the second d-axis voltage command value v d *, the second q-axis voltage command value v q *, and the second f-axis voltage command value v f * by adding the non-interference voltages v d_dcpl , v q_dcpl , and v f_dcpl  calculated in step S 7  to the first d-axis voltage command value v d_dsh , the first q-axis voltage command value v q_dsh , and the first f-axis voltage command value of v f_dsh  calculated in step S 6 , respectively. 
     In Step S 9 , the coordinate conversion unit  410  performs a coordinate conversion for the second d-axis voltage command value v d *, the second q-axis voltage command value v q *, and the second f-axis voltage command value v f * calculated in Step S 8 , thereby calculating the voltage command values v u *, v v *, and v w * for each uvw phase. 
     With the control described above, the controller  2  executes the processes of steps S 1  to S 9  to generate the command values for controlling the motor  4 . Out of the generated command values, the voltage command values v u *, v v *, and v w * calculated in step S 9  are applied to the stator winding of the motor  4  through the PWM converter  102  and the inverter  103 . The second f-axis voltage command value v f *, which is calculated in step S 8 , is applied to the rotor-side winding of the motor  4  through the f-axis current output unit  105 . In this way, the rotation control of the motor  101  is performed. 
     As such, the above explanation for the motor control method for controlling the f-axis current i f , taking into account the f-axis voltage saturation is described above. To apply such oscillation suppression control processing to the motor control, it is necessary to apply processing to the magnetic flux estimator  902  that includes the magnetic flux estimator  802  provided by the oscillation suppression calculation processing unit  2   b  by taking the f-axis voltage saturation into account. 
       FIG. 24  illustrates a control block diagram of the magnetic flux estimator  902  of the second embodiment. The magnetic flux estimator  902  of this embodiment includes control blocks  2401  and  2404 , a multiplier  2402 , a control block  2403 , a limiter  2405 , and an adder  2406 . 
     The control block  2401  represents an f-axis model that models the transfer characteristics from the f-axis voltage v f  to the f-axis current i f . The f-axis model has a characteristic (τ q s+1)/(L f s+R f ). In the control block  2401  takes the f-axis current normative response v fc_lim  by taking into account the f-axis voltage saturation characteristics outputted from the limiter  2405  as an input, and the control block  2401  calculates the f-axis current normative response i f_ref  that by taking into account the transfer characteristics from the f-axis voltage v f  to the f-axis current i f , and outputs the response to the multiplier  2402  and the control block  2404 . 
     The multiplier  2402  calculates the magnetic flux estimate φf{circumflex over ( )} by multiplying the f-axis current normative response i f_ref  by a mutual inductance Mf between the stator and the rotor. The mutual inductance Mf may be obtained by using the value at any operating point of the motor  4  (representative operating point) or by referring to the map data stored in advance. 
     The control block  2403  includes the gain G af . The gain G af  is shown in equation (30) above. The control block  2403  outputs the value obtained by multiplying the input f-axis current command value i f1 * by the gain G af  to the adder  2406 . 
     The control block  2404  represents a filter consisting of the gain G bf  and 1/(τ q s+1). The gain G bf  is shown in Equation (30) above. The control block  2404  outputs the value obtained by the filtering process to the f-axis current normative response if ref to the adder  2406 . 
     The adder  2406  calculates the f-axis voltage command value v fc  by adding together the output values of the control blocks  2403  and  2404  respectively. The calculated f-axis voltage command value v fc  is outputted to the f-axis limiter  2405 . 
     As described above, the magnetic flux estimator  902  of the present embodiment has a control block  2403  and a control block  2404 , and the current F/B system (f-axis current F/B model) is configured by multiplying the f-axis current command value in* by the gain G af  and by multiplying the f-axis current normative response i i_ref  by the gain G bf . This allows the f-axis current response to match the transfer characteristics of the primary delay (see Equation (6)) when there is no f-axis voltage saturation. 
     The f-axis limiter  2405  simulates the f-axis voltage saturation characteristics by limiting the f-axis current command value v fc  according to the supply voltage V dc . This allows the magnetic flux estimator  902  to calculate the f-axis current normative response i i_ref  in which the f-axis voltage saturation characteristic is taken into account in the control block  2401  arranged in the later stage. 
     In addition, the oscillation suppression calculation processing unit  2   b  (see  FIG. 8 ) of this embodiment can calculate the q-axis current normative response i q2 * taking the q-axis current response delay into account by applying the phase advance compensation (τ q s+1) of the q-axis current response to the control blocks  2401  and  2404  provided by the magnetic flux estimator  802 . In other words, the magnetic flux estimate φf{circumflex over ( )} of this embodiment is calculated by performing phase advance compensation for q-axis current response for the f-axis model and the f-axis current F/B model in the pseudo-F/B system composed of the f-axis model modeled to have the characteristics for converting the f-axis voltage v f  to the rotor current i f , the f-axis current F/B model in which the f-axis current command value i f1 * and the output of the f-axis model are inputted, and the f-axis limiter  2405  for limiting the output of the f-axis current F/B model. This allows the oscillation suppression calculation processing unit  2   b  to properly simulate the f-axis current response when there is f-axis voltage saturation. 
     Next, the reluctance torque equivalent magnetic flux estimator  901  of this embodiment is described. When the current is controlled so that the d-axis current response time constant and q-axis current response time constant match, the configuration of the reluctance torque equivalent magnetic flux estimator  901  can be simplified compared to the configuration shown in the first embodiment (see  FIG. 10 ).  FIG. 25  illustrates a control block diagram of the reluctance torque equivalent magnetic flux estimator  901  that is simplified. The reluctance torque equivalent magnetic flux estimator  901  of this embodiment includes a multiplier  2301 . 
     The multiplier  2301  calculates the reluctance torque equivalent magnetic flux estimate φr{circumflex over ( )} by multiplying the d-axis current command value i d1 * outputted from the first current command value calculator  801  by the difference L d −L q  between the d-axis inductance L d  and the q-axis inductance L q . The d-axis inductance L d  and q-axis inductance L q  may be obtained by using the values at any operating point of the motor  4  (representative operating point) or by referring to a map data stored in advance. In the case where the current is controlled so that the d-axis current response time constant and q-axis current response time constant are the same, when the q-axis current response is compensated for phase advance for the transmission characteristics of the primary delay that simulates the d-axis current response delay, the relevant transfer characteristics becomes one. For this reason, the configuration of the reluctance torque equivalent magnetic flux estimator  901  shown in this embodiment can be simplified compared to the configuration shown in the first embodiment (see  FIG. 10 ). 
     By applying the magnetic flux estimator  802  above-mentioned to the oscillation suppression calculation processing unit  2   b , the response delay of the magnetic flux and the effect of the reluctance torque can be taken into account to suppress the drive shaft torsional oscillation. 
     In the following, the effects of the control method (oscillation suppression control process) of the electric vehicle of the second embodiment are described with reference to  FIG. 26 . 
       FIG. 26  illustrates a time chart showing the control results of this embodiment. The horizontal axis represents time, the vertical axis represents motor torque command value [Nm], vehicle front/rear acceleration [m/s2], and f-axis voltage [V] in the order from left, and q-axis current command value [A], d-axis current command value [A], and f-axis current command value [A] in the order from top right. The solid line in the figure shows this embodiment, and the dotted line shows the control using conventional technology (conventional example). 
       FIG. 26  illustrates a scene in which the motor torque command value is changed in a stepwise manner at the timing of t 1  to accelerate (rising) during the vehicle is decelerating by the regenerative torque of the motor  4 . In the control represented in this time chart (this embodiment and the conventional control), the motor torque command value (final torque command value) that suppresses drive shaft torsional oscillation by taking into account the effect of gear backlash is calculated by applying the method disclosed in JP5900609B. 
     In the control of this embodiment in this time chart, the f-axis current normative response time constant (see the control block  701  (τ f )) is set to a value at which f-axis voltage saturation occurs. As shown in the solid line, when this embodiment is applied, the motor torque that suppresses the drive shaft torsional oscillation is realized by the q-axis current command value i q2 * calculated by considering the d-axis current i d  and f-axis current i f  (see  FIG. 8 ), and it can be seen that the vehicle front/rear acceleration oscillation is suppressed. 
     On the other hand, in the conventional controlling method, since the d-axis current and the f-axis current are not taken into consideration, the acceleration oscillation occurs in front and the rear direction of the vehicle due to the effect of the backlash of the gears. 
     According to the control method of the electric vehicle of the second embodiment described above, the magnetic flux estimate φf{circumflex over ( )} of this embodiment is calculated by performing the phase advance compensation in the system that has the f-axis model, which is modeled with the characteristics to convert from the f-axis voltage V f  to the f-axis current i f , and the f-axis limiter  2405 , which limits the output of the f-axis current F/B model. In the phase advance compensation, the q-axis current response is phase advancing compensated against the f-axis model and the f-axis current F/B model. This allows the oscillation suppression calculation processing unit  2   b  to properly simulate the f-axis current response when there is f-axis voltage saturation. 
     The above description of the embodiment of the present invention is only a partial example of the application of the present invention, and is not intended to limit the technical scope of the present invention for the specific configuration of the above embodiment.