Patent Publication Number: US-6903539-B1

Title: Regulated cascode current source with wide output swing

Description:
TECHNICAL FIELD 
   The present invention relates generally to circuit design, and more particularly to a or a current source with enhanced output impedance. 
   BACKGROUND 
   An ideal current source has infinite output impedance and, as a result, provides a constant current over a wide operating voltage range. However, in reality, current sources have finite output impedance and limited output voltage swing. Furthermore, in low voltage applications, a low compliance voltage, Vcompl, may be desired to minimize the output voltage overhead. 
   A commonly used solution to maximize the output impedance involves the use of a regulated cascode current source. The regulated cascode current source offers the desired high output impedance. Another solution involves the use of an operational amplifier to enhance the regulated cascode current source. The use of the operational amplifier reduces the compliance voltage, Vcompl, which can make the design more suitable for low voltage applications. 
   One disadvantage of the prior art is that the regulated cascode current source suffers from a high compliance voltage, Vcompl, the voltage needed to avoid triode region operation. The high value of the compliance voltage, Vcompl, can prevent the use of the regulated cascode current source in low voltage applications. 
   A second disadvantage of the prior art is that the operational amplifier enhanced regulated cascode source requires a high gain operational amplifier, which can increase component count and overall limit on the bandwidth of the current source. 
   SUMMARY OF THE INVENTION 
   These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by preferred embodiments of the present invention which provides a regulated cascode current source with a wide output voltage swing. 
   In accordance with a preferred embodiment of the present invention, a current source comprising a first stage coupled to an input current source, the first stage containing circuitry to receive an input current provided by the input current source, a second stage coupled to the first stage, the second stage comprising a first transistor and a second transistor serially coupled together, wherein a first terminal of the second transistor is coupled to a second terminal of the first transistor, a third transistor having a first terminal coupled to a third terminal of the first transistor, and a level shifter coupled to a third terminal of the third transistor and the first terminal of the second transistor, the level shifter containing circuitry to elevate a voltage at a third terminal of the second transistor, wherein the level shifter is arranged in a source-follower configuration is provided. 
   In accordance with another preferred embodiment of the present invention, a current source comprising a first stage coupled to an input current source, the first stage comprising a first transistor and a second transistor serially coupled together, wherein a first terminal of the second transistor is coupled to a second terminal of the first transistor, a third transistor having a first terminal coupled to a third terminal of the first transistor, a second level shifter coupled to a third terminal of the third transistor and the first terminal of the second transistor, the second level shifter containing circuitry to elevate a voltage at a third terminal of the second transistor, the current source further comprising a second stage coupled to the first stage, the second stage comprising a fourth transistor and a fifth transistor serially coupled together, wherein a first terminal of the fifth transistor is coupled to a second terminal of the fourth transistor, a sixth transistor having a first terminal coupled to a third terminal of the fourth transistor, and a level shifter coupled to a third terminal of the sixth transistor and the first terminal of the fifth transistor, the level shifter containing circuitry to elevate a voltage at a third terminal of the fifth transistor, wherein the level shifter is arranged in a source-follower configuration is provided. 
   In accordance with another preferred embodiment of the present invention, a current source comprising a first stage coupled to an input current source, the first stage containing circuitry to receive an input current provided by the input current source, a second stage coupled to the first stage, the second stage comprising a first transistor and a second transistor serially coupled together, wherein a first terminal of the second transistor is coupled to a second terminal of the first transistor, a level shifter coupled to a third terminal of the second transistor and a second terminal of the first transistor, the level shifter containing circuitry to elevate a voltage at the third terminal of the second transistor, wherein the level shifter is arranged in a source follower configuration, and a third transistor having a third terminal coupled to the level shifter is provided. 
   An advantage of a preferred embodiment of the present invention is that the current source has a high output impedance which provides for a wide output voltage range. 
   A further advantage of a preferred embodiment of the present invention is that the current source has a low compliance voltage, permitting use in low voltage applications. 
   The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures or processes for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
       FIG. 1  is a diagram of a prior art regulated cascode current source; 
       FIG. 2  is a diagram of a prior art regulated cascode current source with an operation amplifier enhancement; 
       FIG. 3  is a diagram of a prior art regulated cascode current source with a pair of level shifters to reduce compliance voltage; 
       FIG. 4  is a diagram of a prior art regulated cascode current source with a level shifter to reduce compliance voltage; 
       FIGS. 5   a  and  5   b  are diagrams of a wide-swing regulated cascode current source with a pair of level shifters in a source-follower configuration, according to a preferred embodiment of the present invention; 
       FIG. 6   a  is a diagram of a wide-swing regulated cascode current source with a pair of level shifters made from P-type MOSFETs in a source-follower configuration, according to a embodiment of the present invention; 
       FIG. 6   b  is a diagram of a wide-swing regulated cascode current source with a pair of level shifters made from N-type MOSFETs in a source-follower configuration, according to a preferred embodiment of the present invention; 
       FIG. 7  is a diagram of a level shifter made from a plurality of P-type MOSFETs, according to a preferred embodiment of the present invention; and 
       FIG. 8  is a data plot of output current versus output voltage for a prior art regulated cascode current source and a wide-swing regulated cascode current source, according to a preferred embodiment of the present invention. 
   

   DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
   The making and using of the presently preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention. 
   The present invention will be described with respect to preferred embodiments in a specific context, namely a regulated cascode current source with a large output voltage swing for use in wireless devices. The invention may also be applied, however, to other applications wherein a large output voltage swing is desirable, along with low voltage considerations. 
   With reference now to  FIG. 1 , there is shown diagram illustrating a prior art regulated cascode current source  100  arranged in a current mirror configuration with a large output impedance. The output (current, I OUT , and voltage, V OUT ) of the regulated cascode current source  100  (or current source) can be regulated by the magnitude of an input current (I IN    105 ). The output voltage, V OUT , of the current source  100  may have a minimum allowable value to prevent the current source from operating in triode region operation and can be expressed as:
 
 V   OUT,MIN   =V   T,N3   +V   DSAT,N3   +V   DSAT,N1   =V compl
 
wherein, V DSAT,N1  is the saturation voltage of transistor MN 1   110 , V DSAT,N3  is the saturation voltage of transistor MN 3   115 , and V T,N3  is the threshold voltage of transistor MN 3   115 . Note that V DSAT,N1  may be expressed as V GS,N1 −V T,N1  of the transistor MN 1   110 , wherein V GS  is the gate-source voltage. The output impedance (R OUT ) of the current source  100  may be approximated with the expression:
 
 R   OUT =( g   m1   *g   m2   *r   ds1   *r   ds2   *r   ds3 )/2
 
wherein, g m1  is the transconductance of transistor MN 1   110 , g m2  is the transconductance of transistor MN 2   120 , r ds1  is the source-drain resistance of transistor MN 1   110 , r ds2  is the source-drain resistance of transistor MN 2   120 , r ds3  is the source-drain resistance of transistor MN 3   115 . Clearly, the output impedance of the current source  100  is large, but the current source  100  may not be suitable for low voltage applications due to its high compliance voltage, Vcompl.
 
   With reference now to  FIG. 2 , there is shown a diagram illustrating a prior art regulated cascode current source  200  with an operational amplifier (op-amp)  205  enhancement to help reduce the compliance voltage, Vcompl. Once again, the output of the current source  200  can be regulated by an input current (I IN    210 ). The op-amp  205  may be used to control the state of a transistor MN 1   215 . For example, the op-amp  205  may compare a voltage against a bias voltage, with the output of the op-amp  205  controlling the voltage at the gate of the transistor MN 1   215 . With the addition of the op-amp  205 , the compliance voltage, Vcompl, can be expressed as:
 
 V compl= V   DSAT,N2   +V   DSAT,N1 
 
wherein, V DSAT,N2  is the saturation voltage of transistor MN 2   220  and V DSAT,N1  is the saturation voltage of transistor MN 1   215 . Hence, the Vcompl of the current source  200  can be lower than the Vcompl of the current source  100  due to the absence of the V T,N3  (from FIG.  1 ). The output impedance (R OUT ) of the current source  200  may be approximated with the expression:
 
 R   OUT   =g   m1   *r   ds1   *r   ds2 *(1+ A )
 
wherein, g m1  is the transconductance of the transistor MN 1   215 , r ds1  is the source-drain resistance of the transistor MN 1   215 , r ds2  is the source-drain resistance of the transistor MN 2   220 , and A is the gain of the op-amp  205 .
 
   Note that a high-gain op-amp  205  may be needed to provide suitable output impedance to the current source. Additionally, the use of a high gain op-amp can increase the component count of the current source  200  and can place a limit upon the bandwidth of the current source  200 . 
   With reference now to  FIGS. 3 and 4 , there are shown diagrams illustrating prior art designs of wide-swing cascode current sources  300  and  400 . The use of level shifters (level shifters  305  and  310  ( FIG. 3 ) and level shifter  405  (FIG.  4 )) can help in reducing the value of the compliance voltage, Vcompl. For both current sources  300  and  400 , the compliance voltage, Vcompl, may be expressed as:
 
 V compl= V   DSAT,N1   +V   DSAT,N2 
 
wherein, V DSAT,N1  is the saturation voltage for transistor MN 1  (transistor  310  ( FIG. 3 ) and transistor  415  (FIG.  4 )) and V DSAT,N2  is the saturation voltage for transistor MN 2  (transistor  315  ( FIG. 3 ) and transistor  420  (FIG.  4 )). For both current sources, output impedance is similar to the output impedance of the current source  100  (R OUT =(g m1 *g m2 *r ds1 *r ds2 *r ds3 )/2). Vcompl can be reduced with the presence of the level shifter as the gate terminal voltage of MN 1  can be biased such that its source terminal voltage can be pushed as low as V DSAT,N1 , before the entire current source goes out of saturation.
 
   However, in the case of the current source  300  (FIG.  3 ), the best current mirror performance may be achieved when an important matching condition is met, the currents I 3  (current source  320 ) and I 4  (current source  325 ) should match. Since I 1 +I 3 =I 4 +I OUT , then I OUT =(I 1 +I 3 )−I 4 . Therefore, in order for I OUT =I 1 , I 3 , should match I 4 . If I 3  and I 4  are poorly matched, the current mirroring accuracy can be impacted significantly. A similar matching situation can be present in the current source  400  (FIG.  4 ). 
   With reference now to  FIG. 5   a,  there is shown a diagram illustrating a wide-swing cascode current source  500 , wherein the current source  500  features a high output impedance and a low compliance voltage, according to a preferred embodiment of the present invention. The current source  500  makes use of a pair of source-follower (S-F) level shifters  505  and  510  to help reduce the compliance voltage, Vcompl. With the use of the S-F level shifters  505  and  510 , the compliance voltage, Vcompl, can be as low as Vcompl=V DSAT,N1 +V DSAT,N2 , wherein V DSAT,N1  is the saturation voltage for transistor MN 1   515  and V DSAT,N2  is the saturation voltage for transistor MN 2   520 . The output impedance of the current source  500  can be similar to the output impedance of the current source  300  (FIG.  3 ), namely, R OUT =(g m1 *g m2 *r ds1 *r ds2 *r ds3 )/2. 
   With reference now to  FIG. 5   b,  there is shown a diagram illustrating a wide-swing cascode current source  550 , wherein the current source  550  features a high output impedance and a low compliance voltage, according to a preferred embodiment of the present invention. The current source  550  can be similar to the current source  500  ( FIG. 5   a ) in that it makes use of a pair of S-F level shifters  555  and  560  to help reduce the compliance voltage, Vcompl. However, rather than using NMOS transistors, the current source  550  makes use of PMOS transistors. 
   With reference now to  FIG. 6   a,  there is shown a diagram illustrating a wide-swing cascode current source  600 , wherein the current source  600  features a high output impedance and a low compliance voltage, according to a preferred embodiment of the present invention. As displayed in  FIG. 6   a,  the S-F level shifters  505  and  510  may be constructed out of current sources  607  and  612  and P-type MOSFET transistors  609  and  614 . Note that the S-F level shifters  505  and  510  are arranged in a source-follower configuration with transistors in the current source  600 . The compliance voltage can be reduced with the help of a level shifter since it allows the drain terminal voltage of transistor MN 2   620  to be lowered and fixed at a certain low voltage such as V DSAT,N2 , which is also the source terminal voltage of transistor MN 1   615 . The normal level shifter configuration in current source  300  ( FIG. 3 ) and  400  ( FIG. 4 ) can pose a problem in I OUT  accuracy as the biasing currents in the level shifters would constitute part of I OUT . Thus, I OUT  accuracy depends heavily on the matching of the level shifters. However, the current source in  500  ( FIG. 5 ) may not play a part in the I OUT  equation. The level shifters  505  and  510  can merely provide the function of proper biasing for the transistors MN 1   615 , MN 2   620 , MN 3   625  and MN 4   630 . In  FIG. 6   a , the current source may be present for NMOS sinking current source configuration. A PMOS sourcing current source  650  can be implemented by using a complementary architecture as shown in  FIG. 6   b.    
   With reference now to  FIG. 7 , there is shown a diagram illustrating a source-follower level shifter  510 , according to a preferred embodiment of the present invention. The S-F level shifter  510 , as displayed in  FIG. 7 , illustrates an alternative preferred embodiment of the present invention. In  FIG. 6   a,  the S-F level shifter  510  was shown with a single P-type MOSFET transistor (transistor MP 1   614 ). However, in certain situations, such as when I BIAS  (current source  705 ) is large, the presence of multiple P-type MOSFET transistors (transistors  710 ) arranged in parallel can sink the large I BIAS . The use of the multiple transistors in parallel can be useful in a low power design. Note that a similar embodiment using N-type MOSFET transistors can be possible with the S-F level shifter  560  ( FIG. 6   b ). 
   With reference now to  FIG. 8 , there is shown a data plot illustrating a comparison of output current versus output voltage for a prior art current source (such as current source  100  displayed in  FIG. 1 ) and for a wide-swing current source (such as current source  600  displayed in  FIG. 6   a ), according to a preferred embodiment of the present invention. A first curve  805  displays the output current versus output voltage for the prior art current source, while a second curve  810  displays the output current versus output voltage for the wide-swing current source. For both curves, above a certain voltage (different for each curve), the output current becomes stable. This voltage is the compliance voltage, Vcompl. For the prior art current source (the first curve  805 ), the compliance voltage is approximately 0.6 volts while for the wide-swing current source (the second curve  810 ), the compliance voltage is approximately 0.2 volts. Since the output currents for both curves level off at approximately the same level, output impedance of the two current sources are similar. 
   Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. 
   Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, composition of matter, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.