Patent Publication Number: US-8988027-B2

Title: Motor control apparatus and motor control method

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2012-181129, filed on Aug. 17, 2012, the entire contents of which are incorporated herein by reference. 
     FIELD 
     The embodiment discussed herein is directed to a motor control apparatus and a motor control method. 
     BACKGROUND 
     Recently, a motor control apparatus has been put to practical use that includes a speed estimator that estimates the rotational speed of a motor from an induced voltage generated by the rotation of the motor and perform sensorless control. In this type of the motor control apparatus, there is known a motor control apparatus that performs maximum torque control in which the reluctance torque is effectively used. The maximum torque control is performed by using a speed estimated error generated by intentionally inducing an error in the true value of a q-axis inductance, which is used by the speed estimator for calculation, and causing the control axis to match the maximum torque operating point. 
     First and second methods are disclosed in Japanese Patent Application Laid-open No. 2009-291072 as a method of shifting the q-axis inductance, which is a parameter for calculation in the speed estimator, from its true value. The first method is a method that uses a parameter L for calculation, which is set to a value satisfying Ld≦L&lt;Lq. The second method is a method that introduces a dm-qm coordinate system, in which the rotational axis, whose direction matches the direction of the current vector that realizes the maximum torque control, is the qm axis and the rotational axis orthogonal to the qm axis is the dm axis, and that uses a parameter Lm for calculation. Motor parameters Ld, Lq, and Φa are used for calculating the parameter Lm for calculation. 
     However, the first method has a problem in that because the parameter L for calculation is a fixed value, the accuracy of the maximum torque control is reduced as the load increases. 
     In the second method, the parameter Lm for calculation is a function of the q-axis inductance; therefore, the second method can improve the first method. However, because the motor parameters (Ld, Lq, and Φa) are used for calculating the parameter Lm for calculation, if there is an error in the initial setting value or the motor parameters change due to temperature or load, an error occurs in the parameter Lm for calculation in accordance with the error in the motor parameters. Therefore, there is a problem in that the maximum torque control cannot be obtained and moreover the speed estimator becomes unstable. 
     SUMMARY 
     A motor control apparatus according to the embodiment includes a current reference generating unit, a current detecting unit, a voltage reference generating unit, a drive unit, a rotational position estimating unit, a change amount estimating unit, and an inductance estimating unit. The current reference generating unit generates a current reference, in which a high frequency signal whose frequency is higher than a drive frequency of a motor is superposed. The current detecting unit detects an output current to the motor from the drive unit. The voltage reference generating unit generates a voltage reference on a basis of a deviation between the current reference and the output current. The drive unit drives the motor on a basis of the voltage reference. The rotational position estimating unit estimates a rotational position of a rotor from a motor parameter including a q-axis inductance of the motor on a basis of the output current and the voltage reference. The change amount estimating unit estimates a change amount of an output torque with respect to a current phase change of the motor corresponding to the high frequency signal. The inductance estimating unit estimates an inductance value that obtains a maximum torque on a basis of the change amount of the output torque with respect to a current phase change and sets the inductance value in the rotational position estimating unit as the q-axis inductance. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
       A more complete appreciation of the invention and many of the attendant advantages thereof will be readily obtained as the same becomes better understood by reference to the following detailed description when considered in connection with the accompanying drawings, wherein: 
         FIG. 1  is a diagram illustrating a configuration of a motor control apparatus according to an embodiment; 
         FIG. 2  is a diagram illustrating a configuration example of a high-frequency current controller; 
         FIG. 3  is a diagram illustrating a configuration of a motor output extracting unit included in a maximum torque controller; 
         FIG. 4  is a diagram illustrating a configuration of a phase-change-amount estimating unit included in the maximum torque controller; 
         FIG. 5  is a diagram illustrating a configuration example of an inductance calculator; 
         FIG. 6  is a diagram illustrating a configuration example of a speed and magnetic pole position estimator; 
         FIG. 7A  is a diagram illustrating a configuration example of a PLL controller; 
         FIG. 7B  is a diagram illustrating another configuration example of the PLL controller; 
         FIG. 8  is a diagram illustrating a flow of a first calculation process of a rotor angular frequency estimated value and a rotor position estimated value; and 
         FIG. 9  is a diagram illustrating a flow of a second calculation process of a rotor angular frequency estimated value and a rotor position estimated value. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Hereinafter, an embodiment of a motor control apparatus and a motor control method disclosed in the present application will be described in detail with reference to the accompanying drawings. This invention is not limited to the following embodiment. 
       FIG. 1  is a diagram illustrating a configuration of a motor control apparatus according to an embodiment. As illustrated in  FIG. 1 , a motor control apparatus  1  according to the embodiment is connected between an AC source  2  and an AC motor  3 . The motor control apparatus  1  extracts power necessary for driving the AC motor  3  from the AC source  2  and supplies it to the AC motor  3 . The AC motor  3  is, for example, an interior permanent magnet synchronous motor (IPMSM). 
     The motor control apparatus  1  includes a power converting unit  10 , an output current detecting unit  11 , and a control unit  12 . The power converting unit  10  includes a converter unit  10   a , an inverter unit  10   b , and a smoothing capacitor C 1 , and supplies the power supplied from the AC source  2  to the AC motor  3  after AC-AC conversion. The power converting unit  10  is an example of the drive unit and the output current detecting unit  11  is an example of the current detecting unit. 
     The converter unit  10   a , for example, includes a rectifier circuit and rectifies the AC voltage supplied from the AC source  2 . The voltage rectified by the converter unit  10   a  is smoothed by the smoothing capacitor C 1  so as to be converted to a DC voltage. The inverter unit  10   b , for example, includes two upper and lower switching elements for each phase of the output phase and supplies the DC voltage output from the converter unit  10   a  to the AC motor  3  after converting it to an AC voltage by the switching elements. The AC motor  3  is driven by the AC voltage output from the inverter unit  10   b . The switching element is, for example, an IGBT (Insulated Gate Bipolar Transistor) or a MOSFET. 
     The output current detecting unit  11  detects the output current flowing to the AC motor  3  from the inverter unit  10   b . Specifically, the output current detecting unit  11  detects instantaneous values Iu, Iv, and Iw (hereinafter, described as output current values Iu, Iv, and Iw) of the output current flowing to the AC motor  3  from the U phase, V phase, and W phase, which are output phases of the inverter unit  10   b , respectively. The output current detecting unit  11  is, for example, a current sensor that detects current by utilizing a Hall element that is a magnetoelectric converting element. 
     The control unit  12  outputs a voltage having desired amplitude and frequency from the inverter unit  10   b  by controlling each switching element of the inverter unit  10   b , thereby driving the AC motor  3 . 
     The control unit  12  includes subtractors  20  and  24 , a speed controller  21 , a injection signal generator  22 , a injection signal coordinate converter  23 , a current controller  25 , a high-frequency current controller  26 , a decoupling controller  27 , an adder  28 , and a PWM calculator  29 . Furthermore, the control unit  12  includes a coordinate converter  30 , a maximum torque controller  31 , an inductance calculator  32 , and a speed and magnetic pole position estimator  33 . 
     The injection signal coordinate converter  23  is an example of the current reference generating unit and the current controller  25 , the high-frequency current controller  26 , the decoupling controller  27 , and the adder  28  are an example of the voltage reference generating unit. Moreover, the maximum torque controller  31  is an example of the change amount estimating unit, the inductance calculator  32  is an example of the inductance estimating unit, and the speed and magnetic pole position estimator  33  is an example of the rotational position estimating unit and the angular frequency estimating unit. 
     The subtractor  20  obtains a deviation between a rotor angular frequency reference ωref and a rotor angular frequency estimated value ωest and outputs it to the speed controller  21 . The rotor angular frequency reference ωref is a reference that defines the angular frequency (hereinafter, described as the rotor angular frequency) of the rotor included in the AC motor  3  and is input from a not-shown upper-level control apparatus. “est” indicates that it is an estimated value. 
     The speed controller  21 , for example, includes a PI (Proportional Integral) controller and generates a δ-axis current reference Iδ_ref by PI control such that the deviation between the rotor angular frequency reference ωref and the rotor angular frequency estimated value ωest becomes zero. The δ-axis current reference Iδ_ref is output from the speed controller  21  to the injection signal coordinate converter  23  and the decoupling controller  27 . 
     The injection signal generator  22  generates a injection signal S mag , which is a high frequency signal, and outputs it to the injection signal coordinate converter  23 . The injection signal S mag  is a signal defined by A mag  sin(fh×2πt). Moreover, the injection signal generator  22  outputs a signal sin(fh×2πt) to the maximum torque controller  31 . “fh” indicates the frequency of the injection signal S mag  and is set to a value higher than the frequency of the voltage that drives the AC motor  3 . 
     Moreover, A mag  is an amplitude of a phase of a current reference vector Is defined by a γ-axis current reference Iγ_ref and the δ-axis current reference Iδ_ref. The frequency fh and the amplitude A mag  of the injection signal S mag  are set not to interfere in consideration of the control response of the speed controller  21  and the switching frequency of the inverter unit  10   b.    
     In the present embodiment, in the rotating coordinate system that rotates at the same speed as the rotational speed of the magnetic flux generated by permanent magnets arranged in the rotor of the AC motor  3 , the direction of the magnetic flux generated by the permanent magnets is defined as the d-axis and the rotational axis for control corresponding to the d-axis is defined as the γ-axis. Moreover, the phase advanced by 90° in an electrical angle from the d-axis is defined as the q-axis and the rotational axis for control corresponding to the q-axis is defined as the δ-axis. 
     The injection signal coordinate converter  23  obtains γδ-axis current references Iγ_href and Iδ_href when the phase of the current reference vector Is is varied by the injection signal S mag  by the following Equation (1) and outputs them to the subtractor  24 . The γ-axis current reference Iγ_ref is, for example, set to zero. 
     
       
         
           
             
               
                 
                   
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     The subtractor  24  subtracts a γ-axis current detected value Iγ_fb to be described later from the γ-axis current reference Iγ_href on which the injection signal S mag  is superposed. Moreover, the subtractor  24  subtracts a δ-axis current detected value Iδ_fb to be described later from the δ-axis current reference Iδ_href on which the injection signal S mag  is superposed. Then, the subtractor  24  outputs each subtraction result to the current controller  25  and the high-frequency current controller  26 . 
     The current controller  25  generates a γ-axis voltage reference Vγ and a δ-axis voltage reference Vδ such that a deviation between the γ-axis current reference Iγ_href and the γ-axis current detected value Iγ_fb and a deviation between the δ-axis current reference Iδ_href and the δ-axis current detected value Iδ_fb each become zero. The current controller  25  is, for example, composed of a PI controller. The current controller  25  outputs the generated γ-axis voltage reference Vγ and δ-axis voltage reference Vδ to the adder  28 . 
     The high-frequency current controller  26  generates γδ-axis voltage references Vγ_href and Vδ_href such that a deviation between the γδ-axis current references and the γδ-axis current detected values becomes zero. The above-described injection signal S mag  is a relatively high-frequency wave; therefore, the high-frequency current controller  26 , which is highly responsive compared with a normal current control, is provided to cause the current value to follow the injection signal S mag . The high-frequency current controller  26  is, for example, composed of a P (proportional) controller, thereby maintaining the stability. 
       FIG. 2  is a diagram illustrating a configuration example of the high-frequency current controller  26 . As illustrated in  FIG. 2 , the high-frequency current controller  26  includes P controllers  40  and  41 . Proportional gains Kp_γ and Kp_δ of the P controller  40  are obtained by the following Equation (2). In the following Equation (2), ω ACR     —     hf  is the configuration parameter and is, for example, set to a value obtained by multiplying the cutoff frequency in the high-frequency current control by 2π.
 
 Kp   —   γ=L   MTPA ×ω ACR     —     hf  
 
 Kp   —   δ=L   MTPA ×ω ACR     —     hf   (2)
 
     The high-frequency current controller  26  updates an inductance value used for calculating the proportional gains Kp_γ and Kp_δ online by an inductance compensation value L MTPA  from the inductance calculator  32 . Consequently, it is possible to reduce the change in the current response due to the setting error of the inductance. In terms of the current controller  25  and the decoupling controller  27 , the parameters are not updated online to maintain the stability. The term online indicates the state where the motor control apparatus  1  is operating. 
     The P controller  40  has the above-described proportional gain Kp_γ and generates the γ-axis voltage reference Vγ_href by proportional control such that a deviation between the γ-axis current reference Iγ_href and the γ-axis current detected value Iγ_fb becomes zero. Moreover, the P controller  41  has the above-described proportional gain Kp_δ and generates the δ-axis voltage reference Vδ_href by proportional control such that a deviation between the δ-axis current reference Iδ_href and the δ-axis current detected value Iδ_fb becomes zero. 
     Returning to  FIG. 1 , an explanation of the control unit  12  is continued. The decoupling controller  27  generates a γ-axis interference voltage Vγ_dcp and a δ-axis interference voltage Vδ_dcp that cancel the effect of the γ-axis current component and the δ-axis current component each interfering with the other current component as the δ-axis voltage and the γ-axis voltage. 
     Specifically, the γ-axis current reference Iγ_ref, the δ-axis current reference Iδ_ref, and the rotor angular frequency estimated value ωest to be described later are input to the decoupling controller  27  and the decoupling controller  27  obtains the γ-axis interference voltage Vγ_dcp and the δ-axis interference voltage Vδ_dcp by the following Equation (3):
 
 Vγ   —   dcp =−ωest× Lq×Iδ _ref
 
 Vδ   —   dcp =ωest× Ld×Iγ _ref  (3)
 
where Lq is the q-axis inductance of the AC motor  3  and Ld is the d-axis inductance of the AC motor  3 . Lq may be set equal to L MTPA  and Ld may be set equal to Ld*, or Lq may be set equal to L MTPA  and Ld may be set equal to L MTPA , on the basis of the inductance compensation value L MTPA  output from the speed and magnetic pole position estimator  33  to be described later.
 
     The adder  28  generates a γ-axis voltage reference Vγ_ref and a δ-axis voltage reference Vδ_ref by adding the output of the decoupling controller  27 , the output of the current controller  25 , and the output of the high-frequency current controller  26 . The γδ-axis voltage references Vγ_ref and Vδ_ref are output to the PWM calculator  29 , the maximum torque controller  31 , and the speed and magnetic pole position estimator  33  from the adder  28 . 
     Specifically, the adder  28  generates the γ-axis voltage reference Vγ_ref by adding the γ-axis interference voltage Vγ_dcp, the γ-axis voltage reference Vγ, and the γ-axis voltage reference Vγ_href. Moreover, the adder  28  generates the δ-axis voltage reference Vδ_ref by adding the δ-axis interference voltage Vδ_dcp, the δ-axis voltage reference Vδ, and the δ-axis voltage reference Vδ_href. 
     The PWM calculator  29  performs rotating coordinate conversion on the γ-axis voltage reference Vγ_ref and the δ-axis voltage reference Vδ_ref by using rotor position estimated value θest and furthermore performs two-phase to three-phase conversion to generate voltage references Vu, Vv, and Vw corresponding to the U phase, V phase, and W phase, respectively. Then, the PWM calculator  29  generates a drive signal that drives the switching elements of the inverter unit  10   b  by a method, such as a triangular wave comparison, on the basis of the voltage references Vu, Vv, and Vw and supplies the drive signal to the inverter unit  10   b . Consequently, the voltage corresponding to the voltage references Vu, Vv, and Vw is output to the AC motor  3  from the inverter unit  10   b.    
     The output current values Iu, Iv, and Iw output from the output current detecting unit  11  are input to the coordinate converter  30  and the coordinate converter  30  performs coordinate conversion to the γ-δ axis coordinate system by using the rotor position estimated value θest after performing two-phase to three-phase conversion on the output current values Iu, Iv, and Iw. The γ-δ axis coordinate system is a rotating coordinate system that rotates in synchronization with the rotor angular frequency estimated value ωest. 
     The coordinate converter  30  obtains the γ-axis current detected value Iγ_fb, which is a γ-axis component, and the δ-axis current detected value Iδ_fb, which is a δ-axis component, by the coordinate conversion to the γ-δ axis coordinate system and outputs them to each of the subtractor  24 , the maximum torque controller  31 , the inductance calculator  32 , and the speed and magnetic pole position estimator  33 . 
     The maximum torque controller  31  obtains a phase change amount Δθ MTPA  on the basis of the γδ-axis current detected values Iγ_fb and Iδ_fb, the γδ-axis voltage references Vγ_ref and Vδ_ref, and a signal sin(fh×2π), which has a frequency and a phase same as those of the injection signal S mag . The phase change amount Δθ MTPA  is a phase change amount of the current reference vector Is after the control is started and is output to the inductance calculator  32 . 
     The maximum torque controller  31  obtains the phase change amount Δθ MTPA  on the basis of a motor input power Pe. Specifically, the γδ-axis current detected values Iγ_fb and Iδ_fb and the γδ-axis voltage references Vγ_ref and Vδ_ref are input to the maximum torque controller  31  and the maximum torque controller  31  obtains the motor input power Pe that is the power input from the power converting unit  10  to the AC motor  3  by the following Equation (4).
 
 Pe=Vγ _ref× Iγ   —   fb+Vδ _ref× Iδ   —   fb   (4)
 
     The motor input power Pe includes a copper loss Pc due to the winding resistance of the AC motor  3  and a component of a reactive power Pr in addition to a motor output power P mecha  that is a mechanical output of the AC motor  3 . The copper loss Pc includes only a DC component. Moreover, the reactive power Pr includes a frequency component whose frequency is the same as that of the injection signal S mag  and a frequency component whose frequency is twice that of the injection signal S mag . The component whose frequency is the same as that of the injection signal S mag  is out of phase by π/2 with respect to the phase of the injection signal S mag . On the other hand, the motor output power P mecha  includes a component whose frequency and phase are the same as those of the injection signal S mag . 
     The maximum torque controller  31  extracts a motor output power fluctuation range Po, which is the amplitude value of a component whose frequency and phase are the same as those of the injection signal S mag , in the motor output power P mecha  from the motor input power Pe by a motor output extracting unit  50  illustrated in  FIG. 3 .  FIG. 3  is a diagram illustrating the configuration of the motor output extracting unit  50  included in the maximum torque controller  31 . 
     As illustrated in  FIG. 3 , the motor output extracting unit  50  includes a band-pass filter (BPF)  51 , a multiplier  52 , and a low-pass filter (LPF)  53 . The BPF  51  is set to pass a signal of the frequency fh and extracts a frequency component P BPF , whose frequency is the same as that of the injection signal S mag , from the input motor input power Pe. 
     The output of the BPF  51  is input to the multiplier  52  and is multiplied by a signal sin(fh×2πt) whose frequency and phase are the same as those of the injection signal S mag . Consequently, the signal, whose frequency and phase are the same as those of the injection signal S mag , in the output of the BPF  51  becomes a DC component and a signal Ph, which includes this DC component, is output from the multiplier  52 . 
     The signal Ph output from the multiplier  52  is input to the LPF  53 , and only the DC component is extracted in the LPF  53  and is output from the LPF  53 . This DC component is a component that corresponds to the motor output power fluctuation range Po. The motor output power fluctuation range Po can be represented by the following Equation (5): 
                   Po   =       3   4     ⁢   ω   ⁢           ⁢   r   ×   Amag   ×   Isa   ×     {             (     Ld   -   Lq     )     ×   Isa   ×                 cos   ⁢     (     2   ⁢   θ   ⁢           ⁢   avg     )       +     λ   ⁢           ⁢   f   ×     cos   ⁡     (     θ   ⁢           ⁢   avg     )                 }               (   5   )               
where ωr is the rotor angular velocity, Isa is the current amplitude of the current reference vector Is, πf is the flux linkage constant, Ld is the d-axis inductance, Lq is the q-axis inductance, and θavg is the phase of the current reference vector Is. In the present embodiment, the γ-axis current reference I_γref is equal to zero; therefore, θavg is the phase of the δ-axis.
 
     On the other hand, a change ∂Te/∂θ of a motor generated torque Te with respect to the phase variation of the current reference vector Is is represented by the following Equation (6): 
                       ∂   Te       ∂   θ       =         3   ⁢   P     4     ×   Amag   ×   Isa   ×     {             (     Ld   -   Lq     )     ×   Isa   ×                 cos   ⁢           ⁢   2   ⁢   θ     +     λ   ⁢           ⁢   f   ×   cos   ⁢           ⁢   θ             }               (   6   )               
where P is the number of motor poles, λf is the flux linkage constant, and Ld and Lq are the d-axis inductance and the q-axis inductance, respectively. Isa is the magnitude of the current reference vector Is and θ is the phase of the current reference vector Is of the AC motor  3 .
 
     Comparing the Equation (5) and the Equation (6), it is found that the motor output power fluctuation range Po is proportional to the change ∂Te/∂θ of the motor generated torque Te with respect to the phase variation of the current reference vector Is. Therefore, the phase of the current reference vector Is in which the motor output power fluctuation range Po becomes zero becomes the maximum torque axis. In the motor control apparatus  1  in the present embodiment, the motor output power fluctuation range Po is estimated as the change ∂Te/∂θ of the output torque Te of the AC motor  3  with respect to the current phase change. 
     The maximum torque controller  31  includes a phase-change-amount estimating unit that detects the phase change amount Δθ MTPA  of the current reference vector Is at which the motor output power fluctuation range Po becomes a setting value Po* of the motor output power fluctuation range. The phase change amount Δθ MTPA  is a phase change amount of the current reference vector Is after the control is started and is a phase change amount that causes the δ-axis to match the maximum torque axis in a steady state. The current phase when the control is started is a phase obtained when the inductance value set in the motor control apparatus  1  as an initial value is used for calculation in an extended electromotive force observer  80  to be described later. The setting value Po* of the motor output power fluctuation range is normally set to zero or near zero; however, it can also be set to other values. 
       FIG. 4  is a diagram illustrating the configuration of a phase-change-amount estimating unit  60  included in the maximum torque controller  31 . As illustrated in  FIG. 4 , the phase-change-amount estimating unit  60  includes a subtractor  61 , a limiter  62 , switches  63  and  64 , a comparator  65 , a PI controller  66 , an adder  67 , an amplifier  68 , and a limiter  69 . 
     The subtractor  61  subtracts the motor output power fluctuation range Po from the setting value Po* of the motor output power fluctuation range and outputs the subtraction result to the limiter  62  and the switch  63 . The limiter  62  is a lower limiter. If the subtraction result of the subtractor  61  is less than zero, the limiter  62  outputs zero to the switch  63 , and, if the subtraction result of the subtractor  61  is equal to or more than zero, the limiter  62  directly outputs the subtraction result of the subtractor  61  to the switch  63 . 
     The switch  63  is controlled by a reference signal S SW  to be described later, which is output from the inductance calculator  32 , and selects and outputs one of the output of the subtractor  61  and the output of the limiter  62  to the switch  64 . Specifically, when the reference signal S SW  is at a low level, the switch  63  selects the output of the subtractor  61  and outputs it to the switch  64 , and, when the reference signal S SW  is at a high level, the switch  63  selects the output of the limiter  62  and outputs it to the switch  64 . 
     If the inductance compensation value L MTPA  reaches the limit value of a limiter  78  (see  FIG. 5 ), the reference signal S SW  is output as a signal at a high level. In this case, if the result obtained by subtracting the motor output power fluctuation range Po from the setting value Po* of the motor output power fluctuation range is negative, zero is input to the PI controller  66  by the limiter  62  and the switch  63 . Therefore, updating of the integrated value in the PI controller  66  is stopped and updating of the phase change amount Δθ MTPA  is stopped. 
     The switch  64  selects one of the output of the switch  63  and the output of the amplifier  68  on the basis of the output of the comparator  65  and outputs it to the PI controller  66 . The comparator  65  controls the switch  64  by comparing a start power P start  with the motor input power Pe. If the motor input power Pe is less than the start power P start , a signal at a low level is output from the comparator  65 . If the motor input power Pe is equal to or more than the start power P start , a signal at a high level is output from the comparator  65 . The comparator  65  may compare the start power P start  with the motor output power fluctuation range Po instead of the motor input power Pe. 
     When a signal at a low level is output from the comparator  65 , the switch  64  selects the output of the amplifier  68 , which inverts the integrated output of the PI controller  66 , and outputs the output of the amplifier  68  to the PI controller  66 . Therefore, in a state where the motor input power Pe is less than the start power P start , the signal, which is obtained by inverting the integrated output of the PI controller  66 , is output to the PI controller  66 . Consequently, the output of the phase-change-amount estimating unit  60  is attenuated or maintained at zero by the set time constant of the PI controller  66 . 
     Therefore, in an area in which the electric energy is small, the estimation operation of the phase change amount Δθ MTPA  by the phase-change-amount estimating unit  60  is stopped and the phase change amount Δθ MTPA  output from the phase-change-amount estimating unit  60  becomes zero or converges to zero. The motor output power fluctuation range Po is calculated from the motor input power Pe; therefore, the motor output power fluctuation range Po is largely affected by the detection accuracy of the output current detecting unit  11  and the output voltage error. Thus, in an area in which the electric energy is small, the accuracy of the motor output power fluctuation range Po degrades. 
     Therefore, in the maximum torque controller  31 , in an area in which the electric energy is small, the operation of the phase-change-amount estimating unit  60  stops. Consequently, the low accuracy phase change amount Δθ MTPA  can be prevented from being output from the phase-change-amount estimating unit  60 . It is desirable that the start power P start  be determined, for example, to a value (for example, about 10% of the motor rated capacity), at which the calculation accuracy of the motor output power fluctuation range Po starts to degrade with the motor rated capacity as a reference. 
     On the other hand, when a signal at a high level is output from the comparator  65 , the switch  64  selects the output of the switch  63 . Therefore, in a state where the motor input power Pe is equal to or more than the start power P start , the output of the switch  63  is output to the PI controller  66 . 
     The PI controller  66  includes an amplifier  45  of a proportional gain Kp, an amplifier  46  of an integral gain Ki, and an integrator  47 . The output of the switch  64  is multiplied by Kp by the amplifier  45  and is output to the adder  67 . Moreover, the output of the switch  64  is multiplied by Ki by the amplifier  46 , is integrated by the integrator  47 , and is output to the adder  67 . 
     The adder  67  adds the output of the amplifier  45  and the output of the integrator  47  and outputs the addition result to the limiter  69 . The limiter  69  limits the output from the phase-change-amount estimating unit  60  within a predetermined range. In other words, if the output of the adder  67  is within the predetermined range, the limiter  69  outputs the output of the adder  67  directly as the phase change amount Δθ MTPA , and, if the output of the adder  67  is out of the predetermined range, the limiter  69  outputs the upper limit or the lower limit of the predetermined range as the phase change amount Δθ MTPA . The output of the integrator  47  is inverted by the amplifier  68  and is output to the switch  64 . 
     Returning to  FIG. 1 , an explanation of the control unit  12  is continued. The inductance calculator  32  illustrated in  FIG. 1  obtains the inductance compensation value L MTPA  from the phase change amount Δθ MTPA  of the current reference vector Is output from the maximum torque controller  31  by the following Equation (7). The following Equation (7) can be derived by modifying the above Equation (5) by using the relationship of θavg=Δθ MTPA +π/2. 
     
       
         
           
             
               
                 
                   
                     L 
                     MTPA 
                   
                   = 
                   
                     
                       
                         λ 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         f 
                         × 
                         sin 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         Δ 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         
                           θ 
                           MTPA 
                         
                       
                       
                         Isa 
                         × 
                         cos 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         2 
                         ⁢ 
                         Δ 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         
                           θ 
                           MTPA 
                         
                       
                     
                     + 
                     
                       L 
                       d 
                       * 
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
     The inductance calculator  32  outputs the inductance compensation value L MTPA  obtained as above to the speed and magnetic pole position estimator  33  and the high-frequency current controller  26 . In the above Equation (7), the d-axis inductance Ld* and the flux linkage constant λf are constants set in the motor control apparatus  1  and are, for example, values determined from offline tuning, in which the motor control apparatus  1  is in a non-operating state, or the information on a motor test report. 
       FIG. 5  is a diagram illustrating a configuration example of the inductance calculator  32 . As illustrated in  FIG. 5 , the inductance calculator  32  includes a sine value calculator  71 , a cosine value calculator  72 , an amplifier  73 , multipliers  74  and  75 , a divider  76 , an adder  77 , the limiter  78 , and a filter  79 . 
     The sine value calculator  71  calculates a sine value of the phase change amount Δθ MTPA . A calculation result sin Δθ MTPA  is multiplied by the flux linkage constant λf by the multiplier  74 . The calculation result of the multiplier  74  is output to the divider  76 . 
     The cosine value calculator  72  calculates a cosine value of the phase change amount Δθ MTPA  doubled by the amplifier  73 . The calculation result cos 2Δθ MTPA  is multiplied by a current amplitude Isa of the current reference vector Is by the multiplier  75 . The calculation result of the multiplier  75  is output to the divider  76 . 
     The divider  76  divides the calculation result of the multiplier  74  by the calculation result of the multiplier  75 . The calculation result of the divider  76  is output to the adder  77  and the d-axis inductance Ld* is added thereto by the adder  77 . The addition result of the adder  77  is output via the limiter  78  and the filter  79 . 
     If the phase change amount Δθ MTPA  of the current reference vector Is after the control is started is zero, the inductance compensation value L MTPA  becomes equal to Ld*. The inductance compensation value L MTPA  is updated on the basis of the phase change amount Δθ MTPA  output from the maximum torque controller  31  and is output to the high-frequency current controller  26  and the speed and magnetic pole position estimator  33 . 
     A lower limit is set in the limiter  78  to prevent overcompensation. When the inductance compensation value L MTPA  reaches the lower limit, the limiter  78  outputs the reference signal S SW  at a high level to the switch  63  as an antiwindup operation. Consequently, the switch  63  of the phase-change-amount estimating unit  60  illustrated in  FIG. 4  is switched and the output of the limiter  62  is output from the switch  63 . 
     The limiter  78  compares the input and output signals of the limiter  78 . If the signals are different from each other, the limiter  78  determines that the inductance compensation value L MTPA  reaches the lower limit. Moreover, the lower limit is, for example, set to a value that is a half of the d-axis inductance Ld* so that the inductance compensation value L MTPA  does not become smaller than the value that is a half of the d-axis inductance Ld*. 
     Returning to  FIG. 1 , an explanation of the control unit  12  is continued. The speed and magnetic pole position estimator  33  detects the rotational speed and the magnetic pole position of the rotor of the AC motor  3 . Specifically, the γδ-axis voltage references Vγ_ref and Vδ_ref, the γδ-axis current detected values Iγ_fb and Iδ_fb, and the inductance compensation value L MTPA  are input to the speed and magnetic pole position estimator  33 , and the speed and magnetic pole position estimator  33  obtains the rotor angular frequency estimated value ωest and the rotor position estimated value θest. The speed and magnetic pole position estimator  33  outputs the rotor angular frequency estimated value ωest to the subtractor  20  and outputs the rotor position estimated value θest to the PWM calculator  29  and the coordinate converter  30 . 
       FIG. 6  is a diagram illustrating a configuration example of the speed and magnetic pole position estimator  33 . As illustrated in  FIG. 6 , the speed and magnetic pole position estimator  33  includes the extended electromotive force observer  80 , a phase error calculator  81 , and a PLL controller  82 . 
     The extended electromotive force observer  80  obtains a γ-axis extended electromotive force estimated value εγ_est and a δ-axis extended electromotive force estimated value εδ_est, for example, by the following Equation (8): 
                       ⅆ     ⅆ   t       ⁡     [           I     γ   ⁢           ⁢   _   ⁢           ⁢   est                 I     δ   ⁢           ⁢   _   ⁢           ⁢   est                 ɛ     γ   ⁢           ⁢   _   ⁢           ⁢   est                 ɛ     δ   ⁢           ⁢   _   ⁢           ⁢   est             ]       =       [           -       R   s       L   d                 ω   est     ⁢       L   q       L   d               -     1     L   d             0               -     ω   est       ⁢       L   q       L   d               -       R   s       L   d             0         -     1     L     d   ⁢                           0       0       0       0           0       0       0       0         ]     ⁢             [           I     γ   ⁢           ⁢   _   ⁢           ⁢   est                 I     δ   ⁢           ⁢   _   ⁢           ⁢   est                 ɛ     γ   ⁢           ⁢   _   ⁢           ⁢   est                 ɛ     δ   ⁢           ⁢   _   ⁢           ⁢   est             ]     +       [           1     L   d           0           0         1     L   d               0       0           0       0         ]     ⁡     [           V     γ   ⁢           ⁢   _   ⁢           ⁢   ref                 V     δ   ⁢           ⁢   _   ⁢           ⁢   ref             ]       +       [           H   1           H   2               H   3           H   4               H   5           H   6               H   7           H   8           ]     ⁡     [             I   γ     -     I     γ   ⁢           ⁢   _   ⁢           ⁢   est                     I   δ     -     I     δ   ⁢           ⁢   _   ⁢           ⁢   est               ]                     (   8   )               
where Rs, Ld, and Lq are motor parameters that are calculation parameters. Rs is the primary resistance. Ld is the d-axis inductance. Lq is the q-axis inductance. H 1  to H 9  are the observer gains. The following Equation (9) can be obtained by developing the above Equation (8) into a discrete system. In the following Equation (9), Ts represents the sampling time.
 
     
       
         
           
             
               
                 
                   
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                           s 
                         
                         ⁡ 
                         
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                                     d 
                                   
                                 
                               
                               
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                         ⁡ 
                         
                           [ 
                           
                             
                               
                                 
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                                   1 
                                 
                               
                               
                                 
                                   H 
                                   2 
                                 
                               
                             
                             
                               
                                 
                                   H 
                                   3 
                                 
                               
                               
                                 
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                                   4 
                                 
                               
                             
                             
                               
                                 
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                                   5 
                                 
                               
                               
                                 
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                       ⁡ 
                       
                         [ 
                         
                           
                             
                               
                                 
                                   
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                                   ⁡ 
                                   
                                     ( 
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                                     ) 
                                   
                                 
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                                       ⁢ 
                                       
                                           
                                       
                                       ⁢ 
                                       _ 
                                       ⁢ 
                                       
                                           
                                       
                                       ⁢ 
                                       est 
                                     
                                   
                                   ⁡ 
                                   
                                     ( 
                                     k 
                                     ) 
                                   
                                 
                               
                             
                           
                           
                             
                               
                                 
                                   
                                     I 
                                     δ 
                                   
                                   ⁡ 
                                   
                                     ( 
                                     k 
                                     ) 
                                   
                                 
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                                       ⁢ 
                                       
                                           
                                       
                                       ⁢ 
                                       _ 
                                       ⁢ 
                                       
                                           
                                       
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                                   ⁡ 
                                   
                                     ( 
                                     k 
                                     ) 
                                   
                                 
                               
                             
                           
                         
                         ] 
                       
                     
                   
                 
               
               
                 
                   ( 
                   9 
                   ) 
                 
               
             
           
         
       
     
     As represented by the above Equation (9), the extended electromotive force observer  80  obtains the γδ-axis extended electromotive force estimated values εδ_est and εδ_est at ([k+1]·Ts) seconds on the basis of the γδ-axis current detected values Iγ_fb and Iδ_fb, γδ-axis current estimated values Iγ_est and Iδ_est, the γδ-axis extended electromotive force estimated values εγ_est and εδ_est, the speed estimated value, and the motor parameters at (k·Ts) seconds. 
     At this time, the extended electromotive force observer  80  obtains the γδ-axis extended electromotive force estimated values εγ_est and εδ_est by using the inductance compensation value L MTPA  obtained by the inductance calculator  32  as the q-axis inductance Lq. The extended electromotive force observer  80  selectively performs a first estimation process and a second estimation process on the basis of the setting from the outside. 
     In the first estimation process, the γδ-axis extended electromotive force estimated values δγ_est and εδ_est are obtained by setting Lq to be equal to L MTPA  and setting Ld to be equal to Ld* in the above Equation (9). On the other hand, in the second estimation process, the γδ-axis extended electromotive force estimated values εγ_est and εδ_est are obtained by setting Lq to be equal to L MTPA  and setting Ld to be equal to L MTPA  in the above Equation (9). 
     The γδ-axis extended electromotive force estimated values εγ_est and εδ_est are input to the phase error calculator  81  from the extended electromotive force observer  80  and the phase error calculator  81  obtains a phase error estimated value Δθest by the following Equation (10) and outputs it to the PLL controller  82 . 
     
       
         
           
             
               
                 
                   
                     Δ 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     
                       θ 
                       est 
                     
                   
                   = 
                   
                     
                       tan 
                       
                         - 
                         1 
                       
                     
                     ⁡ 
                     
                       ( 
                       
                         
                           - 
                           
                             ɛ 
                             
                               γ 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               _ 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               est 
                             
                           
                         
                         
                           ɛ 
                           
                             δ 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             _ 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             est 
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   10 
                   ) 
                 
               
             
           
         
       
     
     As represented by the above Equation (9), the γδ-axis extended electromotive force estimated values εγ_est and εδ_est include a voltage error component generated due to the change of speed, the change in the load state, and the parameter error. On the basis of the γδ-axis extended electromotive force estimated values εγ_est and εδ_est including the voltage error component, the phase error estimated value Δθest is obtained as represented by the above Equation (10). Therefore, it is found that the inductance compensation value L MTPA  obtained by the inductance calculator  32  is reflected in the phase error estimated value Δθest. 
     The PLL controller  82  obtains the rotor position estimated value θest and the rotor angular frequency estimated value ωest on the basis of the phase error estimated value Δθest output from the phase error calculator  81 .  FIG. 7A  and  FIG. 7B  illustrate a configuration example of the PLL controller  82 . 
     In the example illustrated in  FIG. 7A , the PLL controller  82  obtains the rotor position estimated value θest and the rotor angular frequency estimated value ωest by controlling such that the phase error estimated value Δθest estimated from the γδ-axis extended electromotive force estimated values εγ_est and εδ_est becomes zero. Specifically, the PLL controller  82  illustrated in  FIG. 7A  includes a PI controller  91  and an integrator  92 , and obtains the rotor angular frequency estimated value ωest by performing the PI control on the phase error estimated value Δθest by the PI controller  91  and obtains the rotor position estimated value θest by integrating the rotor angular frequency estimated value ωest by the integrator  92 . 
     If there is no parameter error and thus there is no error in the phase error estimated value Δθest, the γ-δ axis coordinates, which are a control coordinate system, can be caused to match the d-q axis coordinates, which are a rotor coordinate system, by controlling the phase error estimated value Δθest to zero. In the present embodiment, parameters do not match true values; therefore, even if the phase error estimated value Δθest is zero in the control, the γ-δ axis coordinates do not match the d-q axis coordinates. The parameters are determined such that the δ-axis matches the maximum torque axis, and, as a result, as in the PLL controller  82  illustrated in  FIG. 7A , the δ-axis in the γ-δ axis coordinates, which are a control coordinate system, is controlled to match the maximum torque axis by controlling the phase error estimated value Δθest to zero. Because the current reference vector Is is always on the δ-axis, the maximum torque control can be performed by causing the δ-axis to match the maximum torque axis. 
     Moreover, the PLL controller  82  may have a configuration illustrated in  FIG. 7B . The PLL controller  82  illustrated in  FIG. 7B  includes a PI controller  93 , a divider  94 , an adder  95 , and an integrator  96 . In the PLL controller  82 , the speed is approximately estimated by dividing the δ-axis extended electromotive force estimated value εδ_est by an induced voltage constant φ by the divider  94 , and the rotor angular frequency estimated value ωest is obtained by adding the estimated speed to the output of the PI controller  93  by the adder  95 . Moreover, the rotor position estimated value θest is obtained by integrating the rotor angular frequency estimated value ωest by the integrator  96 . 
       FIG. 8  and  FIG. 9  are diagrams illustrating the flow of the calculation process of the rotor angular frequency estimated value ωest and the rotor position estimated value θest in the control unit  12 .  FIG. 8  illustrates the flow of the first calculation process in the case where the above-described first estimation process is selected and  FIG. 9  illustrates the flow of the second calculation process in the case where the above-described second estimation process is selected. 
     First, the flow of the first calculation process of the rotor angular frequency estimated value ωest and the rotor position estimated value θest will be described with reference to  FIG. 8 . 
     The maximum torque controller  31  of the control unit  12  compares the start power P start  with the motor input power Pe (Step S 10 ). If the motor input power Pe is equal to or more than the start power P start  (Yes in Step S 10 ), the maximum torque controller  31  sets the target for the PI control to Po-Po* (Step S 11 ). On the other hand, if the motor input power Pe is less than the start power P start  (No in Step S 10 ), the maximum torque controller  31  sets the target for the PI control to an inverse of the integral output (Step S 12 ). 
     The maximum torque controller  31  obtains the phase change amount Δθ MTPA  by performing the PI control on the basis of the setting in Steps S 11  and S 12  (Step S 13 ). Then, the inductance calculator  32  obtains the inductance compensation value L MTPA  on the basis of the phase change amount Δθ MTPA  output from the maximum torque controller  31  (Step S 14 ). 
     The speed and magnetic pole position estimator  33  sets the q-axis inductance Lq and the d-axis inductance Ld that are motor parameters. Specifically, the speed and magnetic pole position estimator  33  sets the inductance compensation value L MTPA  as the q-axis inductance Lq and sets the d-axis inductance Ld*, which is preset, as the d-axis inductance Ld (Step S 15 ). Then, the speed and magnetic pole position estimator  33  obtains the rotor position estimated value θest and the rotor angular frequency estimated value ωest on the basis of the motor parameters set in Step S 15  (Step S 16 ). 
     Next, the flow of the second calculation process of the rotor angular frequency estimated value ωest and the rotor position estimated value θest will be described with reference to  FIG. 9 . In this process, the processes in Steps S 11  to S 14  and S 16  are the same as those in the first calculation process illustrated in  FIG. 8  and the process in Step S 25  is different from that in the first calculation process. 
     In Step S 25 , the speed and magnetic pole position estimator  33  sets the inductance compensation value L MTPA  as the q-axis inductance Lq and sets the inductance compensation value L MTPA  as the d-axis inductance Ld. Then, the speed and magnetic pole position estimator  33  obtains the rotor position estimated value θest and the rotor angular frequency estimated value ωest in accordance with the motor parameters set as above. 
     As described above, the motor control apparatus  1  according to the present embodiment includes the maximum torque controller  31 , the inductance calculator  32 , and the speed and magnetic pole position estimator  33 . The maximum torque controller  31  estimates the motor output power fluctuation range Po of the AC motor  3  corresponding to the injection signal S mag  that is a high frequency signal whose frequency is higher than that of the drive frequency of the AC motor  3 . The inductance calculator  32  estimates the inductance compensation value L MTPA  that obtains the maximum torque on the basis of the motor output power fluctuation range Po and sets it in the speed and magnetic pole position estimator  33  as the q-axis inductance. The speed and magnetic pole position estimator  33  estimates the rotor position estimated value θest, which is the rotational position of the rotor of the AC motor  3 , from the motor parameters that include the q-axis inductance Lq set by the inductance calculator  32  on the basis of the γδ-axis current detected values Iγ_fb and Iδ_fb, which are detected values of the output current to the AC motor  3 , and the γδ-axis voltage references Vγ_ref and Vδ_ref. 
     The motor control apparatus  1  according to the present embodiment can set the q-axis inductance in which an error is intentionally induced from its true value by obtaining the inductance compensation value L MTPA  to be set as the q-axis inductance online without using a fixed value and motor parameters (Ld, Lq, and Φa). Therefore, for example, even if there is an error in the motor parameters or variation in the motor parameters, the speed and magnetic pole position estimator  33  can be stably operated and thus the maximum torque control can be accurately performed. 
     Additional advantages and modifications will readily occur to those skilled in the art. Therefore, the invention in its broader aspects is not limited to the specific details and representative embodiments shown and described herein. Accordingly, various modifications may be made without departing from the spirit or scope of the general inventive concept as defined by the appended claims and their equivalents.