Patent Publication Number: US-6665308-B1

Title: Apparatus and method for equalization in distributed digital data transmission systems

Description:
This is a voluntary divisional of a U.S. patent application entitled “APPARATUS AND METHOD FOR SCDMA DIGITAL DATA TRANSMISSION USING ORTHOGONAL CODES AND A HEAD END MODEM WITH NO TRACKING LOOPS”, Ser. No. 08/895,612, filed Jul. 16, 1997 now U.S. Pat. No. 6,307,868, which was a continuation-in-part of a U.S. patent application entitled “Apparatus And Method For Digital Data Transmission Using Orthogonal Codes”, Ser. No. 08/684,243, filed Jul. 19, 1996, now U.S. Pat. No. 6,356,555, published as PCT publication WO 97/08861 on Mar. 6, 1997, which was a continuation-in-part of a U.S. patent application entitled “Apparatus And Method For Digital Data Transmission Over Video Cable Using Orthogonal Cyclic Codes”, Ser. No. 08/588,650, filed Jan. 19, 1996, now U.S. Pat. No. 5,793,759, which was a continuation-in-part of a U.S. patent application entitled “Apparatus and Method For Establishing Frame Synchronization In Distributed Digital Data Communiation systems”, Ser. No. 08/519,630, filed Aug. 25, 1995, now U.S. Pat. No. 5,768,269, all of which are hereby incorporated by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     The invention pertains to the field of bidirectional passband digital communication systems, and, more particularly to the field of improvements in head end or central office modems to remove the phase locked loops therefrom. 
     Digital data communication systems are well known in the art. Many treatises are available that describe them. Among these treatises are: Dixon, “ Spread Spectrum Systems with Commercial Applications ”, Third Edition, 1994 (Wiley &amp; Sons, New York) ISBN 0 471 59342-7; Stallings “ Data and Computer Communications”,  4th Ed. 1994 (Macmillan Publishing Co., New York) ISBN0-02-415441-5; Lee and Messerschmit, “ Digital Communication,  2d Ed.”, 1994 (Kluwer Academic Publishers, Boston), ISBN 0 7923 9391 0; Haykin, “Communication Systems” Third Edition 1994 (Wiley &amp; Sons) ISBN 0 471 57176-8; Elliott,  Handbook of Digital Signal Processing: Engineering Applications , (Academic Press, Inc. San Diego, 1987), ISBN 0-12-237075-9, all of which are hereby incorporated by reference. Generally, the problem which the invention is an attempt to solve is how to get rid of as many continuous tracking loops as possible in a bidirectional digital data communication system. The reasoning for this can be understood from the following discussion. 
     Digital data distributed communication systems can be baseband systems or passband systems. In baseband systems, the transmission media has the capability of transmitting digital pulses between widely separated transmitter and receiver locations. Passband systems require that the digital data be modulated onto a carrier frequency for transmission over the media. 
     Receivers for digital data passband systems can be either coherent or noncoherent. In coherent systems, the receiver has a local oscillator, usually taking the form of a phase locked loop (PLL) which is part of a continuous tracking loop and is maintained in constant phase lock with the phase and frequency of the carrier on which the received data is modulated. Coherent systems can make use of modulation schemes which alter either the phase, frequency or amplitude or any combination thereof of the carrier in accordance with the information content of the digital data to be transmitted. Incoherent systems do not have the local oscillator at the receiver phase locked to the carrier phase and frequency. In these systems, the designers have chosen to ignore the phase of the received signal at the expense of some degradation of the system performance and throughput. 
     Coherent systems can utilize binary or M-ary amplitude shift keying (ASK), phase shift keying (PSK) or frequency shift keying (FSK), as well as M-ary amplitude phase keying (APK) of which QAM (quadrature amplitude modulation) is a special case. Incoherent systems are limited to binary or M-ary ASK, FSK or differential phase shift keying (DPSK). 
     Coherent systems are higher performance systems because they have an additional degree of freedom for use in the modulation scheme which means more complex constellations of symbol sets can be used and more data bits can be encoded in each symbol in the constellation. This translates to greater throughput. 
     However, coherent systems are more complex since they require additional tracking loop circuitry at the receiver to recover the transmitted carrier and use the information so derived to steer the local oscillator so as to maintain its phase and frequency locked to the phase and frequency of the carrier. Usually the local oscillator being steered in the receiver is a PLL or has a voltage controlled oscillator negative feedback system in it (which is at the heart of almost every tracking loop). Carrier synchronization has been achieved by any one of a number of different ways in the prior art including use of PLLs where the carrier is not suppressed or Mth power tracking loops or Costas tracking Loops where the carrier is suppressed. Mth power and Costas tracking loops also contain voltage controlled oscillators as part of the tracking loop. The problem is that PLLs and negative feedback voltage controlled oscillators in tracking loops can and often do lose lock especially where there is rapid change in phase or frequency caused by conditions in the transmission media. When a PLL or other tracking loop loses lock, the system goes out of synchronization and fails to communicate data—its sole purpose in life. 
     All digital data communication also requires clock synchronization in the receiver to the clock in the transmitter because data is sent during discrete times. These discrete times are variously called chip times, bit times or symbol times in the prior art references. The importance of synchronization of the clock in the receiver to the clock in the transmitter is that in all forms of modulation, the amplitude, phase or frequency of the carrier (or some combination of these) must be sampled during each chip time so as to determine which symbol in the alphabet or code set in use was transmitted during that chip time based upon the phase, amplitude or frequency characteristics of the carrier during that chip time. 
     Receiver clock synchronization can be done on either a long term basis or a short term basis. Short term clock synchronization is called, amazingly enough, asynchronous transmission, but in fact the receiver clock is periodically synchronized to the transmitter clock at the beginning of transmission of each “character”. A character is a collection of 5 to 8 symbols which are transmitted over a very short time (usually the symbols or binary bits that only have two states). The receiver clock resynchronizes during each character at the beginning thereof and need not resynchronize until the next character starts. Asynchronous transmission is cheap and less complex since timing synchronization problems caused by transmission of long uninterrupted streams of bits is avoided by sending the bits one character at a time and requiring synchronization between the receiver clock and transmitter clock only during that character. 
     The problem with asynchronous transmission is the high overhead. Each character of 5 to 8 bits must include a start bit, 1 or 2 stop bits and a parity bit. The start bit is used by the receiver to resynchronize its clock. The overhead of 2-3 bits per character of 5-8 bits makes asynchronous transmission inefficient to transmit large volumes of data. Asynchronous transmission can be extended to sending several characters grouped together with a preamble which is long enough for the receiver to synchronize to transmitted before every group of characters and a tracking loop to maintain the receiver clock in synchronization with the transmitter clock during the transmission of the group of characters. The concepts of the invention are applicable to asynchronous transmission where there is a tracking loop in the remote unit receiver but no tracking loop in the central unit receiver and only a periodic or occasional phase adjustment of the master clock and master carrier phase for use by the central unit receiver. 
     Synchronous transmission is a more efficient way of transmitting data since blocks of symbols or bits can be transmitted without start and stop codes. Sampling by the receiver during the middle of each bit or chip time is accomplished by keeping the receiver clock in synchronization with the transmitter clock. This maintenance of clock synchronization has been done in the prior art in many different ways. For example, a separate clock line can connect the transmitter and receiver, but this is impractical in many situations. A way of avoiding this is to embed the clock information in the data signal transmitted from the transmitter and recover the clock in the receiver. 
     Clock recovery has been done in a number of different ways in the prior art including transmitting the clock along with the data bearing signal in multiplexed form and using appropriate filtering of the modulated waveform to extract the clock. Another method is to use a noncoherent detector to first extract the clock and then processing the noncoherent detector output to recover the carrier. Where clock recovery follows carrier recovery, the clock is recovered from demodulated baseband signals. The early-late gate symbol synchronizer has also been used in the prior art to synchronize the receiver clock to the transmitter clock. This type clock recovery takes advantage of the fact that a matched filter output of a filter matched to a rectangular clock pulse is a triangular waveform which can be sampled early before the peak and late after the peak. By changing the timing of the sampling until the early and late samples have equal amplitude, the peak of the matched filter output signal can be found, and this will have a fixed phase relationship to the clock phase. This information is then used to steer a voltage controlled oscillator in a negative feedback system. Again, complicated circuitry centered around a voltage controlled oscillator is needed to recover the clock. 
     A technique called remote loopback or remote clock has been used in the prior art on, for example T 1  type digital data communication phone lines. This technique is similar to the aspect of the invention involving having the remote unit local clock synchronized to the central unit master clock and using that local clock at the remote unit receiver for the remote unit transmitter. It is also similar to the aspect of the invention of using the central unit master clock, after adjustment in phase to synchronize it to the phase of the received clock from the remote unit transmitter, as the clock signal from the central unit receiver. 
     Since PLLs and tracking loops are not always reliable, and add complication and expense to receivers, it is desirable to be able to get rid of them wherever possible. Thus, a need has arisen for a bidirectional digital communication system where continuous tracking loops in the central unit receiver (or the receiver in the unit having the transmitter which transmits with the master clock and master carrier signals) have been eliminated. 
     SUMMARY OF THE INVENTION 
     A bidirectional digital data communication system according to the teachings of the invention will have: a central unit transmitter with any encoder to receive downstream data, encode it and drive any type of digital passband modulator with the encoder receiving a master clock signal from a master clock oscillator and the modulator receiving a master carrier oscillator; a remote unit receiver which has any compatible detector which receives a local carrier reference signal which is synchronized in frequency and phase to the master carrier signal and which is generated by any form of carrier recovery circuit, with the detector driving a decoder to decode the received data and output it, with the decoder receiving a local clock signal which has been synchronized with the transmitter master clock signal by any clock recovery circuit; a remote unit transmitter having any encoder type for receiving upstream data, encoding it and driving any digital passband modulator, the encoder receiving the local clock reference generated by the remote unit receiver clock recovery circuit and the modulator receiving the local carrier reference signal generated by the remote unit carrier recovery circuit; and a central unit receiver with any compatible coherent detector to detect the signal transmitted from the remote unit transmitter, with the central unit detector using the central unit master carrier from the master carrier oscillator in the transmitter but adjusted in phase to account for propagation delay from the remote unit, and with the decoder using the master clock signal from the central unit transmitter master clock oscillator but adjusted in phase for the propagation delay from the remote unit to the central unit. Thus, the central unit has no phase locked loops or other voltage controlled oscillator circuits for clock recovery or carrier recovery. 
     In the preferred embodiment, the master carrier and master clock are recovered in the RUs and used to transmit data upstream along with preamble data preceding payload data. The preamble data from each RU is used by the central unit transceiver to generate an amplitude and phase correction factor for that RU. The signals from that RU are then demodulated using the CU master carrier and demultiplexed and detected using the CU master clock. Phase and amplitude errors in the detection process caused by latency and channel impairments are eliminated or reduced by using the phase and amplitude correction factors developed for this RU from its preamble data. Thus, there is no need for continuous tracking loops in the CU receiver to recover the clock and carrier used by each RU to transmit its data. This single master carrier and master clock concept and the frame synchronization provided by ranging, and the improved throughput and lower error rates provided by the equalization and power alignment processes taught herein are useful in any form of bidirectional digital data distributed communication system regardless of the form of encoding, multiplexing or modulation used. Examples of the types of multiplexing that can be used in such systems are CDMA, TDMA, inverse Fourier, DMT or any other system where orthogonal signals are used to encode each separate channel of data from a source such as sine and cosine signals etc. 
     In the broadest embodiment of the invention involving no continuous tracking loops in the CU receiver to recover RU clock and carrier, the type of central unit transmitter and modulation scheme is not important nor is it important whether the central unit transmits a single channel of digital data downstream or multiplexes several channels. If the central unit transmitter is a multiplexing transmitter, the type of multiplexing is not important. Likewise, the type of detector used in the remote unit receiver is not important as long as it is compatible with the modulation scheme in use and it is not critical whether the central unit transmitter transmits the master carrier or suppresses it or transmits the master clock separate or embeds it in the data so long as the master clock and carrier phase information get transmitted somehow to the RUs such as embedded in the Barker code of the preferred embodiment. Likewise, the type of carrier recovery and clock recovery circuits used in the remote unit to synchronize the local clock and local carrier oscillators to the master clock and master carrier are not critical. Also, the type of decoder used in the remote unit receiver is not critical so long as it is compatible with the type of encoder used at the central unit transmitter. For the remote unit transmitter, any type of encoder and any type of modulator may be used for the upstream data, and the type of encoding and the type of multiplexing, if any, used for the upstream direction need not be the same as the downstream direction. The clock and carrier signals used by the remote unit transmitter are the same clock and carrier signals used by the remote unit receiver. 
     The central unit receiver can use any type of detector that is compatible with the modulation scheme used by the remote unit transmitter. Likewise, the type of decoder used in the central unit receiver is not critical so long as it is compatible with the remote unit transmitter encoder. The structure and operation of the central unit receiver phase detection and adjustment circuit is not critical to the invention. The only requirement on this circuit is that it be able to occasionally or periodically detect any phase differential between the central unit master carrier and the carrier used to transmit by the remote unit transmitter and detect any phase difference between the central unit master clock and the clock information used to transmit the received data. These phase differences are used by the central unit receiver to occasionally or periodically adjust the phase of the master clock and master carrier to match the phases of the carrier and clock signals used by the remote unit transmitter as received at the central unit receiver. 
     The invention is applicable to both asynchronous and synchronous methods of transmission, although synchronous transmission is much more efficient in terms of overhead consumed per unit of payload data delivered. Use of the invention in asynchronous transmission will be useful in asynchronous systems where tracking loops are used to maintain synchronization of the remote unit receiver local clock during transmission of one or more characters in a group. 
     In the preferred embodiment, the transmitters of the RU use synchronous code division multiplexing (SCDMA). SCDMA is defined as transmission of frames of spread spectrum signals with data from different channels spread using orthogonal pseudorandom spreading codes, said frames being synchronously transmitted from different RUs located at diverse locations such that all frames of corresponding frame number from all RUs arrive at the CU modem with their frame boundaries exactly aligned in time with the frame boundaries of the CU frame of the same frame number. The upstream data is then demultiplexed and decoding by the inverse code transformation that was used in the RU transmitter to spread the spectrum of the data using the orthogonal, pseudorandom spreading codes. 
     According to the most preferred embodiment, there is provided a code division multiplexing multiple access (CDMA) scheme using orthogonal codes to encode multiple channels of digital data for simultaneous transmission over a cable television media which is also carrying frequency division multiplexed cable television programming. Further, in this most preferred embodiment, alignment of multiple subscriber remote units at diverse locations on the cable television media to the same frame alignment reference is used to substantially reduce crosstalk between adjacent codes and allow multiple users to simultaneously share the same cable TV media for auxiliary services other than cable TV programming delivery. The ranging process described herein is useful for any digital communication system which delivers data from physically distributed transmitters to a central location in frames, but in the context of a CDMA system on a cable TV plant, it provides for synchronous CDMA which greatly increases system payload capacity. The use of synchronous CDMA coupled with frequency division multiplexing of upstream and downstream data on different frequencies than the cable TV programming provides a system whereby the entire bandwidth devoted to the digital auxiliary services may be simultaneously shared by multiple users who share a plurality of channels. Any of the known ways of achieving frame alignment may be used to achieve synchronous code division multiple access data transmission. In the preferred embodiment, frame alignment is achieved with the bulk of the processing done by the RUs with the CU only acting in a passive role as a sensor for deciding if a Barker code is in the gap, if there is more than one Barker code in the gap, asking for authentication and providing feedback for all of the above and for fine tuning processing to exactly center each RU&#39;s Barker code in the gap. This ranging process is done by alignment of ranging signals transmitted by remote units to guardbands or gaps between frames. 
     One inventive concept disclosed herein is to achieve high noise immunity by spreading the energy of the transmitted data out over time during transmission, and then compressing the energy again at the receiver to recover the data. Spreading the energy of the transmitted data out over time reduces susceptibility to burst errors and impulse noise. In addition to this spreading concept, the spectral efficiency of the system is enhanced by transmitting multiple separate channels of data over the same media without interference by using separate orthogonal codes to encode the data of each channel so that no interference results when all channels are simultaneously transmitted so long as proper frame alignment is maintained. In this way, the spectral efficiency, i.e., a measure of the amount of data that can be sent from one place to another over a given bandwidth, is enhanced without degradation of the data by crosstalk interference. The orthogonality of the codes used for each data stream, i.e., each channel or conversation, minimizes crosstalk between channels where the system is properly aligned, i.e., synchronized. 
     Using cyclic, orthogonal codes for SCDMA further enhances noise abatement by providing the ability to perform equalization using a subset of these codes. Equalization, as that term is used herein, refers to the process of determining the amount of crosstalk between adjacent codes resulting from minor errors of frame timing alignment and then generating phase and amplitude correction factors which can be used to negate the crosstalk. In the preferred embodiment, the orthogonal codes are cyclic codes. 
     In some species within the genus of the invention, code diversity is used to achieve further noise immunity. It has been found that some orthogonal codes are less immune to narrow band interference and other sources of noise than others. To avoid using such codes to spread the data from the same channel or timeslot all the time, code hopping is used in these preferred species of the inventive genus. Code diversity is achieved in several different ways, but, in the preferred embodiment, each transmitter uses a code shuffler circuit and each receiver uses a code deshuffler circuit. All shuffler and deshuffler circuits receive the same seed and generate the same sequence of pseudorandom numbers therefrom. These pseudorandom numbers are used to generate read pointers to a framer memory and write pointers to a buffer memory. The framer memory is where the information vectors or symbols are stored, and the read pointers generated by the shuffler circuits are used to read the timeslot data, i.e., symbol/information vector elements out in pseudorandom fashion and store them in a buffer in accordance with the write pointers generated by the code hopping shuffler circuit. The information vector elements thus stored in the buffer are used to do the matrix multiplication required by the code division multiplexing scheme. Alternatively, the symbol elements may be read out sequentially from the framer memory and stored pseudorandomly in the buffer. 
     The effect of this synchronous CDMA scheme is to “whiten” the noise sources such that no matter how complex the noise signals, the noise can be effectively managed using conventional error detection and correction bits in a forward error correction scheme. The digital data providing the interactive or bidirectional data communication is sent using a CDMA scheme, but for purposes of synchronization, the CDMA scheme is mixed with a TDMA scheme. More precisely, a guardband or gap which is free of data is added between frames of the CDMA signal. Digital data is transmitted in frames, each frame comprising 3 data symbols and a guardband. The guardband is used for non-data usage such as ranging, alignment and equalization. 
     The synchronous CDMA modulation scheme disclosed herein may be used with any shared transmission media and with any apparatus or method that can get all remote units synchronized to the frame timing of the central unit including the ranging/alignment scheme disclosed herein. Other possible methods of synchronizing to the same frame timing are for all remote units and the central unit to receive the same timing reference signals from some source such as internal atomic clocks or from an external source such as a Global Positioning System satellite from which all remote units and the central unit are effectively equidistant. 
     Likewise, the ranging/alignment scheme disclosed herein is useful for any other modulation scheme which transmits digital data in frames, requires frame synchronization and can insert a guardband between the frames. 
     Some species within the inventive genus use M-ary modulation code division multiplexing. Each remote unit receives a time division multiplexed stream of digital data. Each timeslot contains 9 bits of data. Each 9 bits is stored in a framer memory, and is divided into three tribits, each having 3 bits during readout of the memory. Each of the three symbols transmitted each frame is comprised of 144 of these tribits, one for each timeslot or channel. These tribits are encoded with a 4th bit prior to spreading by the code division multiplexing operation. The 4th bit is added by a Trellis forward error correction encoder to each tribit based upon the three bits of the tribit and based upon the previous state for this timeslot&#39;s data during the last frame. This 4th bit adds sufficient redundancy to enable a Viterbi Decoder in the central unit receiver to make a more error free determination of what data was actually sent in the presence of noise without requesting retransmission. The 4th bit also maps each tribit to a 16 point QAM (quadrature amplitude modulation) constellation by using the first two bits to represent the inphase or I axis amplitude and the last two bits to represent the quadrature or Q axis amplitude. Thus, M-ary modulation is used to achieve greater spectral efficiency. 
     With the system described herein, full 10 megabit/second traffic volume per each 6 MHz channel can be achieved in both the upstream and downstream direction over HFC. Unlike conventional CDMA, SCDMA transmission from transmitters like those disclosed herein stays within 6 MHz bands that do not interfere with or effect other adjacent channels. SCDMA has a number of other advantages over pure FDMA and TDMA systems in terms of capacity, scalability and bandwidth allocation. Standard IS-95 asynchronous Code Division Multiple Access spread spectrum systems are hindered by the capacity constraints of the 5-40 MHz upstream channel and the presence of a large amount of noise, and they often require 30 MHz wide channels which creates channel interference problems with neighboring services in the HFC spectrum. The biggest problem with asynchronous CDMA systems is self-generated noise because the RUs are not aligned with each other thereby losing orthogonality and creating a high degree of mutual interference. The higher self-generated noise raises the noise floor and reduces the capacity. SCDMA system insure that the RUs are in frame synchronization with each other and using orthogonal codes to minimize mutual interference as data is sent upstream. Preferably, SCDMA transmitters are also used to send data downstream. With the system described herein, multiple streams of digital data, each having a 64 kbps throughput can be simultaneously sent over a 6 MHz channel with a total 10 Mbps throughput. Each data stream is Trellis encoded, interleaved and spread over the entire 6 MHz using its own individual spreading code. Use of forward error correction and interleaving increases noise immunity to impulse noise, narrowband interference and Gaussian noise. The Trellis coding adds 4.8 dB coding gain, and interleaving enables withstanding long duration impulse noise of up to 100 microseconds without incurring errors. Use of spread spectrum technology adds another 22 dB processing gain. The combination of techniques yield a total 27 dB interference rejection allowing the system to operate in negative Carrier to Noise Plus Interference Ratio. The SCDMA transmitters are combined with TDMA payload data input streams which makes the system extremely scalable. 
     The high capacity of the SCDMA system disclosed herein is made possible by orthogonality which is made possible by the orthogonality of the spreading codes which is a result of the ranging process and the equalization process. The ranging process assures frame synchronization such that all codes arrive from distributed RUs arrive at the CU at the same time. The ranging process is carried out periodically to account for cable expansion/contraction with changing temperature, but the process is transparent to payload traffic in that it does not slow it down, stop it or cause errors. Re-ranging occurs upons certain error conditions and upon disconnect from the network and each powerup. 
     Equalization is achieved by measuring the channel response from each user to the CU and adjusting a precoder at the RU transmitter to “invert the channel”, i.e., predistort the transmitted signal such that it arrives undistorted at the CU. Power alignment by each RU such that each RU transmission reaches the CU at approximately the same power level also helps to minimize mutual interference. 
     Dynamic bandwidth allocation allows as many 64 kbps streams or channels as necessary to be allocated to a particular service so that high demand applications such as video teleconferencing or high speed internet access can be supported simultaneously with low demand applications like telephony over the same HFC link. Bandwidth allocation is managed at the CU through an activity status table in each RU and the CU that indicates the status of each timeslot and code assignments. The CU updates the RU tables by downstream messages. Bandwidth can be guaranteed upon request while other services with more bursty traffic may contend for the remainder of the total 10 Mbps payload. 
     The advantages over TDMA systems include less need for fast acquisition and correspondingly lower sensitivity to narrowband interference. Further, below a certain SNR, TDMA systems may fail altogether. Contention for certain channels and contention affecting adjacent can cause amplifier overload in TDMA systems and can cause severe throughput and performance problems. FDMA systems where each user gets a narrow upstream frequency slice is very susceptible to narrowband noise which can wipe out an entire channel. FDMA systems often try to avoid this problem with frequency reallocation. This complicates and raises the cost of the system by requiring more intelligence. Throughput is also adversely affected as nothing is sent while frequencies are reallocated. Guardbands between channels waste bandwidth, and frequency misalignment degrades FDMA systems. 
     Any method or apparatus that uses these inventive concepts is within the teachings of the invention and is deemed to be equivalent to the apparatus and methods described herein. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of a bidirectional communication system according to the broadest teachings of the invention. 
     FIG. 2A is a diagram of typical upstream or downstream frame structure showing how each frame is separated from its neighbors by a gap. 
     FIG. 2B is a time diagram illustrating how the general ranging process works. 
     FIG. 3 is a time diagram which illustrates a problem which can occur when the network expands. 
     FIG. 4 is a time diagram which illustrates how the problem illustrated in FIG. 3 can be solved in one embodiment. 
     FIGS. 5A through 5C are a flow chart of a general ranging process according to one embodiment. 
     FIG. 6 is a flow chart illustrating one embodiment of how re-synchronization to frame boundaries can be achieved by dead reckoning after the CU changes its delay vector. 
     FIG. 7 is a flow chart illustrating another embodiment of how re-synchronization to frame boundaries can be achieved by a downstream instruction from the CU after the CU changes its delay vector. 
     FIG. 8 is a block diagram of an RU modem according to one embodiment of the invention. 
     FIG. 9 is a block diagram of a framer circuit for use in the CU and RU transmitter sections. 
     FIG. 10 is a timing diagram illustrating the relationships of various clock signals in one embodiment of the system. 
     FIG. 11 is a block diagram of a timebase PLL circuit for use in generating the master clock in the CU or recovering the master clock from a clock steering signal from the frame detector in the RU. 
     FIG. 12 is a diagram of the timing offset relationship between the receiver frame counter and the transmit frame counter in the RUs for purposes of achieving frame synchronization. 
     FIG. 13 is a diagram of how the transmit frame timing delay translates to the state of fill of memory in the framer circuit. 
     FIG. 14 is a diagram illustrating the interleaving of data in the framer circuit and how the framer circuit is emptied for transmission. 
     FIG. 15 is a diagram illustrating the timing relationships between the read and write pointers in the framer circuit. 
     FIG. 16 is a diagram illustrating how tribits of interleaved data are stored as symbols in the framer memory. 
     FIG. 17 is a diagram of the preferred trellis encoder. 
     FIG. 18 is a diagram of the preferred QAM16 constellation. 
     FIG. 19 is a table of the binary and polar representations of each of the QAM16 constellation points. 
     FIG. 20A is a diagram illustrating the matrix multiplication of the information vector for each symbol by the orthogonal code matrix to achieve code division multiplexing. 
     FIG. 20B illustrates the matrix multiplication of the real part of the information vector by the code matrix to generate the real part of the result vector which is sent to the modulator. 
     FIG. 21 is the mapping of the constellation for the fallback mode LSBs. 
     FIG. 22 is a table of the LSB and MSB fallback mode mappings. 
     FIG. 23 is a diagram of one form of modulator that can be used to modulate the spread spectrum data onto two RF carriers. 
     FIG. 24 is a diagram illustrating the problem with rapidly changing chip amplitudes which can cause excessive high frequency content. 
     FIG. 25 is a block diagram of one embodiment for carrier recovery circuitry in the RU receivers to recover the master carrier from pilot channel data and slicer error signals. 
     FIG. 26 is a block diagram of another form of carrier recovery circuitry in the RU receivers to recover the master carrier from pilot channel data and slicer error signals. 
     FIG. 27 is a flow chart of the process carried out by the CU receiver to adjust the phase of the master clock and master carrier signals used in the CU to recover each RU&#39;s data using amplitude and phase error adjustments developed and stored in memory for each RU from preamble data sent by that RU. 
     FIG. 28 is a block diagram of one embodiment of a CU modem. 
     FIG. 29 is a block diagram of one embodiment for a demodulator for used in the CU or RU receivers. 
     FIG. 30 is a more detailed block diagram of one embodiment for an RU receiver. 
     FIG. 31 is a more detailed block diagram of one embodiment for a CU receiver. 
     FIG. 32 is a block diagram for an SCDMA embodiment of a CU transmitter. 
     FIG. 33 is a block diagram for an SCDMA embodiment of an RU transmitter. 
     FIG. 34 is a block diagram for a frame detector/ranging detector useful in the RU and CU receivers to detect Barker codes, do clock recovery etc. 
     FIG. 35 is a timing diagram of the gap acquisition process in the RUs. 
     FIG. 36 is a diagram of the early-late gating process to recover the master clock phase information in the RU receivers. 
     FIG. 37 is a diagram illustrating the data patterns which are acceptable for a centered Barker code to be declared in the fine tuning process. 
     FIG. 38 is a block diagram of one form of implementation of code diversity shuffling. 
     FIG. 39 is an alternative embodiment for a code diversity shuffling circuit. 
     FIG. 40 is another alternative embodiment for a code diversity shuffling circuit. 
     FIG. 41 is another embodiment for a code diversity shuffling circuit. 
     FIG. 42 is a block diagram of a carrierless shaping filter modulator. 
     FIG. 43 is a diagram of the Fourier spectra of the real and imaginary parts of the orthogonally code division multiplexed data. 
     FIG. 44 is a spectrum diagram of the real and imaginary Fourier components of the spread spectrum data after is passed through the shaping filters  1134  and  1136  of the carrierless Hilbert transform modulator  507 . 
     FIG. 45 is a flow chart of a simple, non-boundless RU ranging process. 
     FIG. 46 is a flow chart of a CU side authentication process which counts pulses. 
     FIG. 47 is a flow chart of the process carried out by the CU for a simple, non-boundless ranging. 
     FIG. 48 is a flow chart for the RU side binary stack contention resolution process. 
     FIG. 49 is a flow chart for an RU side ranging and contention resolution process using a binary tree algorithm. 
     FIG. 50 is a diagram of the structure of FFE/DFE filter  764 . 
     FIG. 51 is a block diagram of the kiloframe detector in the frame detector for recovering kiloframe markers from the pilot channel data. 
     FIG. 52 is a diagram of the state machine that monitors frame synchronization. 
     FIG. 53 is a flow diagram of one embodiment of a time alignment, power alignment, upstream and downstream equalization training processes. 
     FIG. 54 is a block digarm of a system using any multiplexing and modulation form for transmisssions downstream and SCDMA RU transmitters for upstream data and uses an RU receiver with no tracking loops for carrier and clock synchronization with the RUs. 
     FIG. 55 is a block diagram of a simple CU SCDMA receiver with no tracking loops for clock and carrier synchronization with the RUs. 
     FIG. 56 is a block diagram of a simple RU SCDMA, FFT −1  or DMT transmitter. 
     FIG. 57 is a block diagram of a simple bidirectional digital data communication system which uses TDMA or any other multiplexing scheme for downstream transmission and synchronous TDMA for upstream transmission. 
     FIG. 58 shows a diagram of the ranging registers as a function of timing offset. 
     FIG. 59 is a simple block diagram of the hardware involved in the equalization structure of the RUs used in the equalization training process of the preferred embodiment. 
     FIG. 60 is a flow chart of the preferred 2-step initial equalization training process. 
     FIG. 61 is a flow chart of the preferred equalization training stability check process. 
     FIG. 62 is a flow diagram of the preferred periodic 2-step equalization training process. 
     FIG. 63 is a flow diagram of the preferred rotational amplifier correction process to insure that the rotational amplifier has not falsely locked on a local minima. 
     FIG. 64 is a flow diagram of the preferred equalization convergence check. 
     FIG. 65 is a flow diagram of the preferred power alignment process. 
     FIG. 66 is a network diagram showing a typical installation of a distributed system wherein the teachings of the invention are useful. 
     FIG. 67 is a diagram showing how the offset register affect the frame number count in the transmitter of RUs in boundless ranging systems. 
     FIG. 68 is a ranging timing diagram for an alternative form of ranging. 
     FIG. 69 is a ranging timing diagram for a distributed system having a maximum TTA of 3 frames. 
     FIG. 70 illustrates a 6 chip wide ranging listening window in a gap. 
     FIG. 71 illustrates how 6 contention vectors V 1  through V 6  are generated from the 6 chips of the listening window of 32 consecutive frames to find contentions or valid IDs. 
     FIG. 72 is a table showing 8 valid IDs arranged on the 8 chips of an 8 chip listening window of 33 consecutive frames, and showing no contentions. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED AND ALTERNATIVE EMBODIMENTS 
     Referring to FIG. 1, there is shown a block diagram of a bidirectional, digital, passband communication system employing the teachings of the invention. The circuits to the left of dotted line  10  represent the central unit modem or transceiver, while circuits to the right of dotted line  10  represent the remote unit modem or transceiver. A message source  12  provides a downstream message signal on line  14  which may be either digital or analog. This signal is received and encoded by an encoder  16  which may be any type of encoder/multiplexer known in the prior art or later developed. The encoder  16  receives a master clock signal on bus  22  from master clock oscillator  24 . The master clock signal on line  22  will be the clock signal to which the entire system synchronizes. The master clock signal defines the bit times or chip times during which a single bit or symbol comprised of multiple bits are used by modulator  20  to modulate the phase, amplitude or frequency of the master carrier signal which the modulator receives on line from a master carrier synthesizer  28 . The function of the encoder, among other things, is to assemble bits from the message source  12  into the groups which define each symbol which is to be transmitted during each chip time defined by the master clock (where a symbol could be a single binary bit), to assemble the chips into frames and to generate the framing signal which marks the frame boundaries. If a multiplexed system exists, the encoder  16  assembles the data to be transmitted on each virtual channel and prepares the data for multiplexing. In the case of code division multiple access or time division multiple access systems, the encoder  16  does the actual multiplexing of data from different sources onto different codes or into different timeslots on bus  18  and generates the frame boundary markers. If a frequency division multiplexed system is in use, the encoder assembles the different data streams to be transmitted on the different frequencies and outputs them on separate buses represented by line  18  to the modulator  20  and generates frame boundary markers. The modulator  20  receives multiple carriers, all phase coherent with the master carrier from the master carrier synthesizer  28 , and modulates each data stream onto its own dedicated carrier. 
     If the signal on line  14  from the message source  12  is analog, one of the functions of the encoder  16  is to sample and digitize it prior to assembly of the digital data into symbols or chips to be transmitted during each chip or bit time. 
     If message source  12 , is digital, it emits one symbol every T seconds with the symbols belonging to an alphabet of M symbols comprised of symbols m 1 , m 2  . . . In binary systems, there are only two symbols, logic 1 and logic 0. In larger alphabets, each symbol may be represented by multiple bits so the output on line  14  is M-ary meaning there are M possible symbols in the alphabet, each of which can be made up of multiple bits. Generally, encoder  16  serves to produce a signal vector S made up of N elements with one such set for each of the M symbols in the source alphabet. 
     The signal vector S is passed on bus  18  to a passband modulator  20 . The details of construction and operation of the modulator  20 , like the details of the construction and operation of the encoder are not critical to the invention, and they can be anything known in the prior art or later developed. The function of the passband modulator  20  is to construct a distinct signal s i (t) of duration T for each symbol m i  by modulating the master carrier signal during each time T using the bits of the symbol to guide the process of altering the phase, frequency or amplitude or some combination thereof in accordance with the selected modulation scheme. 
     Typically, the master clock oscillator  24  is a temperature compensated, crystal controlled oscillator and the clock signal thereof is fed to a master carrier frequency synthesizer  28 , as symbolized by line  30 , for use in synthesis of the master carrier signal. In alternative embodiments, the master carrier synthesizer can be a stand alone crystal controlled oscillator. In other alternative embodiments, the master clock signal need not be generated in the central unit transmitter, and, instead, the master clock signal is received from an external source such as the message source  12 . Likewise, the master carrier can be received from an external source. In still other alternative embodiments, the master clock and master carrier synthesizer need not be crystal controlled and may even vary in frequency so long as they vary slowly enough that the tracking loops in the remote unit receiver can stay in lock. If variable sources for clock and carrier are used, the periodicity of resynchronization process carried out by a phase detect and adjust circuit  32  in the central unit receiver should be made to have a smaller period so resynchronization is done more frequently. More detail on the function of circuit  32  will be given later. 
     After modulation, the signal is transmitted on transmission media  40  to the remote unit or units. Transmission media can be a hybrid fiber coax cable plant, a cellular phone system, a landline telephone network or a local area or wide area network medium connecting computers and peripherals together. Transmission from the central unit to the remote units will be referred to as the downstream direction, while transmission from the remote units to the central unit will be referred to as the upstream direction. The carriers used for upstream and downstream transmission are usually in the RF range of frequencies and are separated in frequency although alternatives discussed below to separate upstream and downstream data may also be used. 
     The downstream RF signal is received by both a coherent detector  44  and a carrier and clock recovery circuit  42 . Carrier and clock recovery circuit  42  is typically a phase locked loop or other type tracking loop. The functions of circuit  42  are: to generate a local clock signal and a local carrier signal internally, usually employing voltage controlled oscillators; detect the phase difference between the received master clock signal and the local clock signal and lock the local clock signal in phase and frequency with the received master clock signal and output the phase and frequency locked local clock signal on line  46 ; and detect the phase difference between the received master carrier signal and the local carrier signal and lock the phase and frequency of the local carrier signal to the phase and frequency of the received master carrier signal and output the phase and frequency locked local carrier signal on line  48 . The manner in which the master carrier and clock signals are transmitted downstream are not critical to the invention, and any of the ways known in the prior art or later developed will suffice the practice the invention. For example, in multiplexed systems, one code, timeslot or frequency may be devoted solely to sending the master carrier or master clock or both. In the particular example of the invention in a synchronous code division multiplexed environment, the master carrier is sent as a pilot tone on a dedicated code, and the clock information is embedded in a unique Barker code which is transmitted during every guardband between frames of payload data. The details of the construction and operation of PLLs and tracking loops are well known in the art, and any of the prior art configurations that are compatible with the particular manner in which the master carrier and master clock signals are transmitted downstream will suffice to practice the invention. Examples of types of tracking loops other than PLLs that are also known in the prior art for use in carrier synchronization and tracking the phase and frequency of a master carrier are Mth power loops and Costas loops described at page 564-565 of the Haykin treatise incorporated by reference herein. The clock synchronization circuitry in circuit  42  can be as simple as an appropriate filter when the master clock is transmitted with the data bearing signal in multiplexed form. Another approach used in the prior art is to use a noncoherent detector to extract the clock signal embedded in the data. Clock recovery can be performed after carrier recovery to recover the clock from baseband signals output from the coherent detector. The preferred method of clock recovery is to use the early-late gating method to sample the output of a matched filter which has a transfer function matched to a particular signal bearing the clock information such as a unique Barker code or a rectangular clock pulse. Some of the prior art clock synchronization techniques are described at pages 566-7 of the Haykin treatise incorporated by reference herein. 
     The coherent demodulator/detector  44  functions to demodulate and detect the incoming symbols and can be any prior art design which is compatible with the modulation scheme used in the central unit transmitter. Although, the detector  44  is stated here to be a coherent detector, in embodiments such as the SCDMA (synchronous code division multiple access) example described below where a rotational amplifier is used to correct phase errors, the demodulator/detector  44  does not have to be coherent. In such embodiments, the demodulator portion does not have to receive a reference carrier which is phase locked with the master carrier so long as the rotational amplifier is used in the detector to correct the resulting phase errors. The same is true for the coherent detector  70  in the CU receiver. 
     The function of the demodulator/detector is to use the local carrier reference signal on line  48  to demodulate the payload data from the downstream RF signal, detect the transmitted constellation points and output a baseband signal on line  50 . 
     The baseband signal is received and processed by any compatible decoder represented by block  52 . The function of decoder  52  is to reverse the encoding and/or multiplexing process carried out by encoder  16  at the central unit transmitter and determine which symbol was sent during every chip time or bit time. The decoder  52  receives the local clock signal on line  46  and uses it to determine when the bit time or chip time boundaries are for purposes of sampling. The decoder  52  also functions to detect the frame boundaries of the downstream frames and reorganize the received data back into the frames organized by the encoder  16  for output on bus  54 . To assist the decoder is doing the frame boundary recognition, a framing signal is generated on line  55  by a frame detector circuit  57 . Any prior art decoder design that can perform this function for the particular encoding/multiplexing scheme selected for use by the central unit transmitter will suffice to practice the invention. The frame detector circuit  57  receives the downstream RF signal on line  59  (or is coupled to the baseband signal output from the detector  44 ) and looks for unique frame boundary signals in the stream of data transmitted from the central unit transmitter. Frame detectors are well known in the art, and there is such a circuit in every digital communication system that transmits data in frames. One method of frame detection used in the SCDMA examples presented below is separation of frames by a guardband, and transmission of a unique Barker code by the central unit transmitter during every guardband. This stream of incoming data at the remote receiver is passed through a filter having a transfer function matched to said Barker code and the correlation peak which results when the Barker code passes through the matched filter is used to mark the frame boundaries. 
     The upstream payload data is received from any message source  56  on line  58 . An encoder  60  receives this message signal (and digitizes it if necessary) and functions like encoder  16  to assemble the bits into symbols for transmission during each bit time or chip time defined by the local clock reference signal on line  46 . As was the case for encoder  16 , encoder  60  may also do the multiplexing of different channels of data onto different codes or into different timeslots in CDMA or TDMA systems, respectively. In the case of an FDMA system, encoder  60  assembles the message bits from one or more sources into separate bits streams which are supplied to a modulator  62  for use in modulating separate carriers received by the modulator from a synthesizer (not shown) which receives the local carrier reference signal on line  48  and generates a plurality of different carriers therefrom. 
     The encoder  60  may use a different form of encoding and/or multiplexing for the upstream direction than were used for the downstream direction. Any encoder design known in the prior art or later developed will suffice for purposes of practicing the invention. In fact, the encoder  60  can even use a clock signal at a different frequency from the master clock signal so long as the different clock signal is phase coherent with the master clock signal. Phase coherent means that there is periodic coincidence in time of clock edges between the two different clock frequencies. In such an embodiment, the local clock reference signal on line  46  (locked in phase and frequency to the master clock) would be supplied to a frequency synthesizer which would then generate the new clock frequency so as to be phase coherent with the master clock signal. 
     The upstream data output from the encoder  60  on line  64  is received by an adjustable delay circuit  65  which receives an adjustable delay value Td. This circuit is used when the system of FIG. 1 requires frame synchronization and the remote units are at differing distances from the central unit. Typically such systems include synchronous TDMA and synchronous CDMA systems. The value of Td is adjusted for each remote unit based upon its physical distance from the central unit so as to achieve frame synchronization. Frame boundaries are delineated by an easily found signal. In frame synchronous systems such as SCDMA, the frame detector  68  can be eliminated. 
     The output of the delay circuit  65  is sent to a modulator  62 . Modulator  62  functions to guide modulation of the phase, frequency or amplitude of some combination thereof of one or more carriers. In a single carrier system, the carrier being modulated is the local carrier reference signal on line  48  which is locked in frequency and phase with the master carrier. In an alternative embodiment, the local carrier reference signal is supplied to a frequency synthesizer which generates a different frequency carrier which is phase coherent with master carrier signal. Phase coherent in this context is that there is periodic coincidence in time of zero crossings of the master carrier and the new carrier frequency generated by the synthesizer. 
     The particular structure and operation for the passband modulator  62  is not critical to the invention. Also, the particular modulation scheme used is not critical to the invention and need not be the same modulation scheme used in the downstream direction. Any prior art design for a modulator or a design subsequently developed will suffice to practice the invention so long as it is compatible with the type of encoding done by encoder  60 . 
     Typically, the transmission in the upstream direction is done at a different frequency from the downstream transmission so as to share the transmission media  40  by frequency division multiplexing. However, other forms of multiplexing such as time division or code division multiplexing may also be used to separate the upstream and downstream data. Frequency translator  66  represents the circuitry needed to separate the downstream and upstream data and it assumes that the form of separation is FDM. If TDMA or CDMA is used to separate the upstream from the downstream data, the circuit  66  represents whatever circuitry is used to do the multiplexing. Such circuitry is well known in the art. 
     The upstream RF signal is transmitted across media  40  to a central unit receiver. The upstream RF is coupled in the central unit receiver to a phase detect and adjust circuit  32 , a frame detector  68  and a coherent detector  70 . The function of the phase detect and adjust circuit is: to occasionally or periodically extract the received clock signal and the received carrier signal from the upstream RF signal; determine the phase difference between the extracted clock and carrier signals and the master clock and master carrier signals, respectively; adjust the phase of the master clock and master carrier signals and apply the phase adjusted clock and carrier signals to the decoder  72  and coherent detector  70 , respectively. 
     The design of the phase detect and adjust circuit  32  is not critical to the invention, and any circuit that can perform the function stated above will suffice to practice the invention. One example of a phase detect and adjust circuit would be a pair of delay lines through which the master clock and master carrier signals are transmitted, with the amount of delay set to equal the total turnaround time for transmission of the carrier and clock signals from the central unit to the remote unit and back to the central unit. If the total turnaround time is stable, this circuit will adjust the phase of the master clock and master carrier signals for phase coherence with the received clock and carrier signals and once the phase is adjusted, it does not have to be adjusted again. If the total turnaround time changes because of, for example, network expansion, the phase detect portion of the circuit can periodically or occasionally determines the phase differences in any one of a number of different ways known in the prior art. For example, phase differences can be determined by comparing the phase between preamble data and Barker code data encoding the carrier and clock data, respectively, said preamble data and Barker code data being transmitted by each RU occasionally or periodically such as at the beginning of each frame or on a dedicated code or in a dedicated timeslot. The phase information recovered from the Barker code and preamble data transmitted by the RU is compared to the phase of the master clock and carrier. 
     In other words, one way of transmitting the clock and carrier information for each RU to the CU is for the remote unit transmitter to generate a Barker code which is a copy of the CU Barker code but which encodes the local clock reference and which has good correlation properties such that it can be detected in the presence of noise or even if transmitted in the midst of payload data. This Barker code can be modulated onto the local carrier reference at the remote unit and transmitted to the CU. This Barker code can be transmitted in guardbands between upstream data frames where it is detected by the phase detect and adjust circuit and used to extract the phase of the local clock and local carrier signals used for upstream transmission. The extracted clock and carrier phase information is compared in phase to the master clock and master carrier phase in the CU for each RU, and the phase error for each RU&#39;s local clock and local carrier reference signals is applied to master clock and master carrier rotational amplifiers, respectively, when data from that RU is being received. These rotational amplifiers are coupled to receive the master clock and master carrier signals, respectively, and to receive the phase error signals and adjust the phase of each signal in accordance with their respective phase errors. The phase adjusted master clock and master carrier signals at the output of the rotational amplifiers are then applied to the decoder  72  and coherent detector  70  via lines  74  and  76 , respectively. If no guardbands are used, the Barker codes for the upstream channel can be transmitted occasionally or periodically during the payload data, and the same process described above is carried out following extraction of the clock and carrier from the Barker code. 
     In alternative embodiments, only the clock is transmitted with the Barker code, and the remote unit carrier is transmitted as a separate sidetone pilot channel on a dedicated code, a dedicated timeslot or on its own frequency which is different than either of the upstream or downstream frequencies. 
     Each remote unit generates its own framing signals. These unique signals are detected by frame detector  68 . The RU framing signals can be Barker codes transmitted by each RU indicating the start of its frame. The details of the design of the frame detector are not critical to the invention, and any frame detector from the prior art or later developed will suffice to practice the invention. A detailed design for a frame detector to detect a Barker code is presented later herein. 
     After the master carrier signal has had its phase adjusted to match the phase at the central unit of the carrier transmitted by remote unit, the phase adjusted carrier on line  76  is used by coherent detector  70  to detect the upstream RF and convert it to baseband observed data on line  80 . Detector  70  can be any coherent prior art detector which is compatible with the form of modulation employed on the upstream channel. The details of the detector design are not critical to the invention. 
     The baseband data output from the detector on line  80  is supplied to decoder  72 . The function of the decoder  72 , like the decoder  52 , is to make a decision from the observed signal as to which symbol was transmitted during each bit time or chip time, and reverse the encoding process performed by encoder  60  by descrambling, de-interleaving and/or demultiplexing the data and reassembling it into the frames of data originally put together by the encoder  60 . This process is done with the aid of the phase adjusted master clock signal on line  74  which defines the boundaries of the bit times or chip times and with the aid of the framing signal on line  82  which defines the frame boundaries. The reassembled data stream is output on line  84 . 
     SPECIFIC EXAMPLES 
     There follows some examples of specific systems that utilize the teachings of the invention. First an embodiment that uses synchronous code division multiple access is discussed. Synchronous code division multiple access in a distributed system requires that all frames from the remote units at different distributed locations arrive at the central unit receiver with their frame boundaries aligned in time. Accordingly, we start with a discussion of the ranging process which is carried out between each remote unit (hereafter RU) and the central unit (hereafter CU) so as to set a transmit frame timing delay for the remote unit which will result in proper frame alignment of that remote unit&#39;s frames. 
     Those skilled in the art will appreciate that the ranging processes described herein has applicability to any distributed data communication system which transmits data in frames regardless of the form of multiplexing or the form of modulation used. Likewise, the equalization processes and power alignment processes described herein all have applicability to any distributed digital data communication system having a near-far problem and having channel impairments that can cause phase and/or amplitude errors in the received points causing them to deviate on the constellation plane at the receiver end from their positions on the same constellation plane at the transmitter end. 
     In the SCDMA system to be described below, the upstream and downstream data is received at the transmitter in a time division multiple access (TDMA) stream with 8 payload bits in each timeslot. The digital data in the TDMA streams is re-arranged into symbols, as described briefly above, and is transmitted in frames, with three symbols plus one guard band or gap per frame. The guardband or gap is reserved for transmission of alignment Barker codes, and no other data is supposed to be transmitted during the gaps. 
     The concept in alignment is to adjust variable delays imposed at the site of each transmitter prior to transmission of a Barker code so as to compensate for different propagation delays from each transmitter site such that the Barker code from each subscriber transmitter trying to align arrives at the head end receiver during the same gap. When the variable delays at each subscriber transmitter are adjusted properly, each subscriber will be said to be in alignment so that the signals encoding the symbols that are of simultaneously transmitted on the transmission media will all be transmitted with the same frame timing. 
     Alignment is important to obtain pure orthogonality so as to obtain low cross talk. If the transmitters are not perfectly aligned, the signals transmitted can still be recovered, but there is some cross talk between channels which will limit the capacity of the SCDMA system to carry information. 
     This process of aligning all the delay circuits in the transmitters is sometimes alternatively called ranging herein and is broadly applicable to other types of multiple access digital data transmission systems also which suffer from different propagation times from different transmitter sites such as time division multiple access systems that form part of the prior art discussed above. 
     Referring to FIG. 2A, there is shown a diagram of the typical frame structure. In the preferred embodiment, each frame is composed of three symbols of 144 chips each and a gap or guardband comprised of 16 chips for a total of 448 chips each having 278 nanoseconds duration. The chip is the basic unit of time in the “code domain”, where code domain refers to the signals propagating across the shared media. In the preferred embodiment, each chip is a QAM modulated element of a result vector where the result vector is comprised of a number of elements equal to the number of timeslots and is the result of code division spreading of the elements of an information vector constructed from the bits of each channel or timeslot. In the preferred embodiment, each receiver receives a TDMA serial bit stream comprised of 144 individual timeslots or channels each of which contains 8 bits. To these 8 bits there is added a 9th bit in the preferred embodiment which can be used for side channel conversations with the CU unrelated to the data received from the external device. These 9 bits are divided into three tribits of 3 bits apiece. A collection of 144 of these tribits is stored in a framer memory and, in some species within the inventive genus, these 144 tribits will be the information vector which is multiplied by the code matrix to generate a result vector having 144 elements. These 144 result vector elements will be QAM modulated to generate the 144 chips that are transmitted as a symbol. This process is repeated for each of the three tribits of each timeslot thereby resulting in the transmission of three symbols in each frame. In the preferred embodiment however, each tribit is encoded with one or more redundant bits based upon the three bits and the state of these same three bits of the same timeslot during the last frame for purposes of forward error correction. The redundant bit(s) is calculated to aid a Viterbi Decoder in a receiver in the central unit to ascertain with a higher degree of accuracy from the received signals which have been corrupted by media impairments what bits were originally present as each tribit. Some species within the inventive genus may omit the addition of the redundant bits and the Viterbi Decoder and still many advantages within the genus of the invention will still be present although a higher bit error rate will result. 
     One skilled in the art will appreciate that the construction of the information vector which will be used to generate each symbol by taking only some of the bits from each timeslot spreads the data from each timeslot out over time. This renders the data less susceptible to burst noise. The code division multiplexing allows multiple channels of digital data to be simultaneously transmitted in a 6 mHz channel without interference between channels. In addition, frequency division multiplexing may be utilized to transmit even more channels of digital data above and beyond the 144 channels transmitted in the first 6 mHz channel. In other words, another 144 different TDMA digital channels may be code division multiplexed and transmitted simultaneously with the first 144 digital channels but on a second 6 mHz channel. This second 6 mHz channel has a different center frequency than the first 6 mHz channel which is separated from the center frequency of the first 6 mHz channel sufficiently to not interfere therewith. Both the first and second 6 mHz channels have center frequencies which are separated sufficiently from the center frequencies of the cable television programming sharing the same media so as to not interfere therewith. In alternative embodiments, this scheme can be replicated with any number of symbols greater than 1, or with only one symbol if immunity to burst noise is not important. 
     In FIG. 2A, the three symbols of frame F n  are symbolized by blocks  162 ,  164 , and  166 . The gap or guardband is symbolized by blocks  161  and  171  on both ends of the frame. There is one guardband associated with each frame. The guardband  171  (sometimes also referred to herein as the gap) is used for synchronization and equalization purposes for the frame comprised of symbols  162 ,  164 ,  166  and guardband  171 . The symbols carry the information for the various channels of digital data provided to the subscribers. The frame period is 125 microseconds. The frame data payload is 128 channels times 72 kilobits per second per channel plus control and management channels each of which has a data rate of 72 kilobits per second for management and control information. 
     The process of synchronization is the process wherein each RU has a variable delay in its transmitter set using feedback from the CU on one of the management and control channels such that the transmitted frame from each RU arrives at the CU with its frame boundaries exactly aligned with the frame boundaries of the frames from the other RUs. Alignment of all frames from all RUs results in the beginning of the gap  161  for each frame from each RU occurring at the same time at the location of the CU regardless of differences in propagation delays from the various RUs to the CU. In FIG. 2A, time increases to the right. 
     Alignment of Any Digital Data System That Sends Data Bits Collected As Frames 
     Referring to FIG. 2B, there is shown a symbolic diagram illustrating the concepts involved in alignment. In FIG. 2B points having increasing positive coordinates along the y-axis starting from the origin at  99  represent increasing time. Points along the x-axis to the right of origin represent increasing distance from the central unit which is designated at position  170 . Time  99  represents the beginning of symbol  162  in FIG. 2A at the CU. The gap  171  at the end of the three symbols will be used for alignment, and the end of gap  171  will be deemed the end of the frame. 
     The alignment process is started asynchronously by any RU that needs to align. The central unit transmits a Barker code during each frame at the same time in the frame. In the preferred embodiment, this Barker code is transmitted during the gap. This Barker code is received by each remote unit at a different time because of different propagation delays, but as to any particular RU, the Barker code is always received at the same time during every frame until the CU changes its delay (a concept to be discussed more fully below). The Barker code represents a trigger to any RU attempting to align and marks the receive frame timing reference for that RU. The time of receipt of the Barker code represents the start of the variable delay interval being adjusted by the RU during the alignment process. 
     The CU&#39;s “every frame” Barker code transmission during the frame shown in FIG. 2A is represented by line  180 . The Barker code is received by RU # 1  at position  167  at time  172 . The Barker code is received by RU # 2  at position  169  at time  174 . The alignment process is a trial and error process of adjusting a delay from the time of receipt of the Barker code to the time of transmission of the same Barker code by each RU back toward the central unit at position  170  until the delay is properly adjusted such that the re-transmitted Barker code arrives at the CU during the gap. Vector  168  represents correct delay timing for RU # 1  at position  167  such that its Barker code transmission  173  (preferably, the RU Barker code is identical to the CU Barker code) arrives in the middle of the gap  171 . Dashed vector  176  represents an incorrect delay resulting in a Barker code transmission, represented by dashed line  178 , from RU # 1  which arrives sometime during the middle of symbol  166  thereby missing the gap  171 . This condition represents an incorrect alignment and may result in crosstalk. 
     Likewise, the RU # 2  at position  169  uses zero delay and emits a Barker code transmission  182  immediately upon receipt of the Barker code trigger transmission  180  from the CU  170 . This Barker code transmission  182  from RU #  2  also arrives during the middle of gap  171  thereby indicating that RU #  1  and RU #  2  are correctly aligned. 
     The alignment Barker code transmissions are typically short bursts having energy levels which are sufficient to make detection during gap  171  easy even though gap  171  also includes random noise energy and with good correlation properties and amplitudes not so high as to substantially interfere with data if the Barker code arrives at the CU in the middle of a symbol. 
     The alignment Barker code transmissions are detected during the gap by performing a correlation mathematical operation in the CU receiver between the Barker code that was transmitted and the received signal. If the received signal was the same Barker code that was transmitted by the CU, the correlation operation will output a signal that peaks at the time of maximum overlap between the Barker code transmitted by the CU and the received signal. The timing of this peak indicates the alignment state of the RU that transmitted the Barker code which resulted in the peak. Each symbol encoded in the code domain includes error detection and correction bits (ECC bits) such that any errors that occur can usually be detected and corrected when the symbols are re-constituted by the framer circuitry in the receiver. 
     Referring to FIG. 3, there is shown a diagram like that of FIG. 2B which illustrates a problem which occurs when the network physically expands. This can occur under certain circumstances such as during the heat of a summer afternoon when the physical media thermally expands thereby altering the propagation times of Barker code signals from the CU to the RUs and from the RUs back to the CU. In the example shown, the CU at position  170  transmits Barker code  196  at time  99 . This Barker code reaches the nearest RU, RU # 1 , at position  190  at time  172 . The same Barker code reaches the furthest RU, RU # 128 , located at position  192  at time  102 . RU # 1  uses a delay symbolized by vector  198  and re-transmits the Barker code  108  at time  138 . This alignment transmission hits gap  106  in frame # 1  indicating that RU # 1  is properly aligned. 
     The RU # 128 , when located at position  192  uses no delay and immediately retransmits Barker code transmission  109  at time  102 . Transmission  109  also arrives during gap  106  indicating that, at least at position  92 , RU # 128  is properly aligned. 
     Now suppose that the network physically expands such that RU # 128  finds itself physically at position  193 . In this position, RU # 128  receives Barker code transmission  196  from the CU at time  103 , and, because RU # 128  is already using the minimum possible delay for retransmission of an alignment code, alignment transmission  110  is also transmitted at time  103 . However, because of the physical expansion of the network, alignment transmission  110  reaches the CU at time  111  which is after the end of the gap  106  and sometime in the middle of the first symbol of frame # 2 . 
     When an RU properly hits the gap, it is authenticated, i.e., identified, and the CU tells it that alignment has been achieved thereby causing the RU to stop adjusting its delay by trial and error. Because RU # 128  does not receive any acknowledgement from the CU that it is properly aligned, its starts incrementing its delay vector in a trial and error process. After several incrementations, the delay vector finally reaches the delay represented by vector  112 . With this delay vector, an alignment transmission  114  is transmitted from RU # 128  at time  113  which reaches gap  116  located at the end of frame  2 . However, this means that RU # 128  is synchronized with the wrong frame. It is required for proper operation of the system to have all RUs synchronized to the gap at the end of the same frame in which the Barker code transmission from the CU which triggered the RUs alignment transmissions occurred. If one or more RU aligns to the gap at the end of another frame, the results can be disastrous in terms of errors generated in the CU receiver in interpreting data transmitted by the RUs. 
     Referring to FIG. 4, there is shown a diagram like that of FIG. 3 which illustrates the solution to this misalignment problem outlined in the discussion of FIG.  3 . In the diagram of FIG. 4, CU  170  imposes a delay, represented by vector  116 , prior to transmitting the alignment triggering Barker code transmission  196  at time  99 . The Barker code transmission  196  arrives at the nearest RU, RU # 1 , at position  190  at time  118 . Time  118  establishes the receive frame timing for RU # 1 . RU # 1  then imposes a delay represented by vector  122  and transmits the same Barker code alignment transmission  124  at time  123 . Time  123  establishes the transmit frame timing reference for RU # 1 . The time delay between times  118  and time  123  is predictable since the CU will transmit its Barker code transmission  196  at the same time during every frame (in the gap) until such time as it is necessary to alter the timing of transmission  196  to keep all RUs in alignment. In other words, the time of reception of the Barker code transmission  196  for all RUs is predictable and will be a periodic signal which happens once during each fame. The alignment transmission  124  from RU # 1  reaches gap  106  at the end of frame # 1 . 
     The alignment transmission  196  from the CU reaches RU # 128 , the furthest RU, at time  120 . Time  120  establishes the receive frame timing reference for RU # 128  while at position  192 . Thereafter, at time  125 , the RU # 128  transmits alignment transmission  128 . This transmission arrives during the gap  106  at the end of the first frame thereby indicating that RU # 128  is properly aligned at this position. 
     As in the case of RU # 1 , the delay between times  120  and  125  for RU # 128  is predictable. 
     Now suppose that the network expands, and RU # 128  finds itself at position  194 . In this position, the CU alignment triggering transmission  196  arrives at time  127 . In order to stay aligned, RU # 128  will reduce its delay vector  126  to zero and immediately retransmit an alignment transmission  130  comprising the same Barker code which it received. The transmission  130  arrives during gap  106  thereby indicating that RU # 128  is still aligned at its new position by cutting its delay vector to zero. 
     Now assume that the network further expands such that RU #  128  finds itself at position  196 . In this new position, alignment transmission  196  from the CU would arrive at time  129 . With a zero delay by RU # 128 , the resulting alignment transmission  131  would arrive at time  133  just after the end of the gap  106  thereby indicating the RU # 128  had been taken out of alignment by the expansion of the network. RU # 128  would then continue to adjust its delay vector until it aligned to the next gap following the end of frame # 2  thereby causing errors. 
     To prevent this from happening, when the CU finds that an RU which was previously in alignment has gone out of alignment because of network expansion, the CU will reduce its initial delay from the delay represented by vector  116  to the delay represented by vector  132 . With this new delay vector, a Barker code alignment triggering transmission  135  will be transmitted at time  137 . This alignment triggering transmission  135  will arrive at the position of RU # 1  at time  139  and will establish a new receive frame timing reference. If RU # 1  has not adjusted its delay vector  122  in advance by one of the mechanisms to be described below, it will go out of alignment. It may then enter a realignment phase and will ultimately, by trial and error, adjust its delay vector to that represented by dashed vector  136 . After so adjusting its delay, RU # 1  will transmit an alignment transmission  124  at time  123  so as to again hit gap  106  thereby re-entering alignment. 
     The alignment triggering transmission  135  from the CU arrives at the position  196  of RU # 128  at time  141 . Using a zero delay vector, RU # 128  transmits its alignment transmission  134 . This alignment transmission  134  arrives during gap  106  thereby placing RU # 128  again in alignment. 
     FIG. 4 shows an alignment process where the alignment is to the gap at the end of the first frame in which the alignment trigger signal  196  is transmitted. In real life systems, this may not be practical, so the alignment process is carried out to the gap following some integer number of frames in the future. The mathematical expression which defines this relationship is given in equation (1) below: 
     
       
           TTA=T   cu   +T   ru +2 ×T   p =constant= n×T   F   (1) 
       
     
     where 
     TTA=the total turnaround time from the CU to the farthest RU; 
     T cu =the delay imposed by the CU illustrated by vector  116  in FIG. 4; 
     T ru =the delay imposed by the farthest RU illustrated by vector  126  in FIG. 4 (also called T far ); 
     2×T p =two times the propagation delay T p  from the CU to the farthest RU; and 
     n×T F =an integer multiple of the frame interval T F . 
     Of course, when the network expands, there is a certain additional delay in the propagation delays which will be called T u  for the uncertainty of this additional propagation delay. Therefore, three additional requirements are imposed with respect to how much delay the CU and the RUs must be able to impose. Those additional requirements are given below in equations (2), (3) and (4): 
     
       
           T   cu   =[T   d   +T   u ]modulo  T   F   (2) 
       
     
     where 
     T d =the span of the network, i.e., equal to the quantity [TTA 2 −TTA 1 ] where TTA 2  equals the total turnaround propagation time for a signal to propagate from the CU to the farthest RU and back, and TTA 1  equals the total turnaround propagation time for a signal to propagate from the CU to the nearest RU and back; and 
     modulo T F =the remainder of [T d +T u ] divided by T F . 
       T   far   &gt;T   u   (3) 
     where 
     T far =the smallest possible T ru  of the farthest RU and is equal to the smallest RU delay which can be imposed by the farthest RU; 
     
       
           T   near   &lt;T   F   −T   u   (4) 
       
     
     where 
     T near =the maximum possible T ru  of the nearest RU. 
     What all this means in a practical sense is that to set up the delays in the network so that all RUs are aligned, the following steps are taken and the limitations on possible delays imposed by the CU and RUs given in equations (1) through (3) are imposed so that all RUs align to the same gap. The practical network to be aligned by the following procedure has a CU coupled by a fiber optic trunk line to an optical node. The optical node is located out in the area to be served and can be coupled to as many as 2000 homes by 2000 individual coaxial links. To align such a network, step 1 would be to bring an RU to the position of the optical node and fix its delay at T near =T F −T u . With this delay, the nearest RU would not hit any gap except by shear luck. Assuming the nearest RU does not hit the gap with this delay, the second step would be to adjust the delay of the CU until the nearest RU hits a gap. When this occurs, the condition T cu =[T d +T u ] modulo T F  would be true meaning that the CU would have adequately compensated for the uncertainty of the propagation delay increment to T d  caused by network expansion. 
     Referring to FIGS. 5A,  5 B, and  5 C, there is shown a flow chart for the general alignment/ranging process which is used in training all RUs to set their transmit frame timing delays T d  properly such that each frame transmitted by an RU will arrive at the CU at the same time as all other frames transmitted from other RUs despite differing propagation times. One of the unique characteristics of the ranging processes described herein is that the RU does the ranging process and the CU is more or less passive which is in contrast with the prior art. 
     How the RUs Synchronize Their Local Oscillators To the Master Carrier and Master Chip Clock Signals From the CU In An SCDMA Embodiment 
     Generally at the time of powerup of an RU, the RU first adjusts its AGC level to make full use of its analog to digital converter dynamic range. Next, the RU does frame detection to determine where the gaps in the CU broadcasts are in time by performing correlations in the RU receiver frame detector looking for the known Barker code which the CU transmits during every gap. Once the gap is located, the frame detector sets the time base generator to synchronize on that receive frame timing reference. Next, the RU performs chip clock synchronization and carrier recovery in the manner described below. Carrier recovery is done by examining slicer error on a known BPSK pilot carrier or pilot channel signal transmitted during a predetermined timeslot using a predetermined code (CU local oscillator signal samples in timeslot  0  spread with all 1 s CDMA code and transmitted using BPSK in the preferred embodiment). 
     In this particular SCDMA embodiment, the pilot channel is the manner in which the master carrier signal from the CU is transmitted to the RUs so that they can synchronize their local oscillator PLLs to the master carrier for purposes of generating their local carrier reference signals. These local carrier reference signals are used by the RU receiver to detect the incoming downstream data and by the RU transmitter to transmit the upstream data. The pilot channel also carries the frame number data. In other words, the RU receiver slicer error on the pilot channel signal is used to synchronize the RU local oscillator to the phase of the CU master carrier local oscillator or other master carrier source. 
     Chip clock synchronization in the RUs to the chip clock, i.e., the master clock signal of the CU, is performed by the fine tuning circuitry of the frame detector in each RU. The frame detector in each RU synchronizes the RU chip clock to the master chip clock signal embedded in the Barker code sent by the CU during every gap. This is all the RU needs to do to set itself up for reception of CU data and messages. 
     The RU then starts listening to CU messages to determine if it tuned to the right CU and to determine when the CU solicits ranging activity by a message on one of the command and control channel. In some embodiments, the “clear to range” message can be eliminated, and the CU can watch for ranging Barker codes all the time, but it is preferred to allow the CU to throttle ranging activity. The RU then performs a ranging process described below and registers itself with the CU by sending an authentication sequence of Barker codes after frame synchronization has been achieved (discussed below). This is done by a CPU in the RU when it receives a message from the CU saying “I found one Barker code in the gap, please send your authentication code”. The RU CPU then sends data to the RU transmitter telling it what authentication sequence of Barker codes to send back to the CU to identify this particular RU. The CU will then transmit a message indicating what authentication code it found and how many chips off center of the gap the Barker code it found landed. The CPU in the RU that is ranging then properly adjusts the transmit frame timing delay reference T d  to center the Barker code in the gap. Other items of data the RU CPU sends to the ranging circuit in the RU transmitter is data indicating the power level to use for the ranging Barker codes and an RU/CU signal indicating to the ranging circuit whether it should follow the rules of ranging for an RU or CU. 
     The CU next instructs the RU to entering an equalization training interval to determine the coefficients to set into the RU transmitter&#39;s precode filter to predistort the RU signals to eliminate channel distortion and test the quality of the ranging result. The training algorithm is discussed below, but other ways of performing equalization which are known in the prior art can also be used. In addition, other ways of achieving frame synchronization known in the prior art can also be used and other ways of achieving synchronization of the RU local carrier oscillator and local clock oscillator to the master carrier and master chip clock signals, respectively, known in the prior art can also be used to practice the invention of eliminating tracking loops in the CU in the SCDMA environment. 
     FIGS. 5A through 5C gives the details of interaction between the CU and RUs to achieve frame synchronization using the particular ranging process symbolized by FIG.  4 . The ranging process starts as symbolized at block  181  with the CU waiting for a predetermined interval from the start of each frame and then sending a trigger signal Barker code transmission to the RUs during the gap. Usually this trigger signal is sent during the gaps between frames even when the CU adds additional delay for reasons discussed below. The RUs monitor these gaps for these Barker codes using their frame detector circuits. 
     Block  183  symbolizes the process wherein each RU trying to synchronize (the terms “synchronize”, “ranging” and “alignment” all are used synonymously to mean the process of training an RU to set its delay vector properly to get its frame boundaries aligned with the CU frame boundaries) receives the Barker code trigger signal transmission from the CU using its frame detector and sets its receive frame timing and then sets a first trial and error delay value for its delay vector. Thereafter, the RU transmits the same Barker code it received from the CU towards the CU as an alignment transmission using the first trial and error delay value. 
     In block  185 , the CU monitors the gap for receipt of a Barker code by performing a correlation between any received signal during the gap and the Barker code that was transmitted as the trigger signal. If a Barker code identical to the trigger signal is received during the gap, the correlation will result in a correlation peak being found in the gap. If a correlation peak is found, processing proceeds to the process symbolized by block  191 . There, the CU broadcasts a message to all RUs indicating that it found activity in the gap. Then the process of block  192  is performed where each RU trying to synchronize sends its “signature”, i.e., its RU identification code in the form of a Barker code transmission sequence. That is, in response to the broadcast from the CU indicating activity in the gap, each RU trying to synchronize sends its unique signature towards the CU in order to determine if that RU&#39;s Barker code is the Barker code the CU found in the gap and whether it is the only RU in the gap. This process is called authentication. 
     The process of block  193  symbolizes the start of the authentication process. Each RU has a unique signature which comprises the transmission and nontransmission of Barker codes during the gaps of a multiple frame authentication period. Specifically, the unique signature of each RU will involve transmitting the Barker code during some gaps of the authentication period but not during others in a sort of “Morse code”. Each Barker code transmission results in a correlation peak during one of the chips in the gap. Each RU has a unique 16 bit RU ID, each bit being either the presence or absence of a Barker code correlation peak somewhere in the gap. Therefore, it takes 16 frames or 4 suprerframes to transmit the RU ID. 
     The number of gaps during which the Barker code is transmitted compared to the number of gaps during which the Barker code is not transmitted during the authentication period is such that if only one RU is aligned to the gap and is transmitting its authentication signature, activity will be found in the gaps of the authentication interval only 50% of the time. This scheme for authentication is chosen so that the CU can detect contentions, i.e., more than one RU in the same gap, in the manner described below. 
     After performing the process of block  193 , the process of block  195  on FIG. 5B is performed. This process involves the CU monitoring each of the gaps during the plurality of signature sequence frames in the authentication interval and performing correlations between the signals received in each of the gaps and the Barker code that the CU transmitted. Correlation peaks are found comparing the correlator output to a threshold value. The threshold value is set by detecting a noise threshold when the gap is empty and setting the threshold at a fixed delta above the empty gap base noise value. 
     Next, the process of block  197  is performed. In this process, the CU counts the number of gaps in the authentication interval that have had activity detected therein, and then compares that number to the total number frames in the authentication interval to determine if the 50% activity level limit has been exceeded indicating that more than one RU is hitting the gap. The advantage of this method is that activity detection, contention detection and authentication are all combined into a single process thereby speeding up the process by more efficiency. 
     Returning to the consideration of the process of block  185 , if the CU, while monitoring the alignment gap for activity, finds no peak resulted from the correlation calculation, then the process of block  186  is performed. In this process, the CU broadcasts a message to all RUs telling them to adjust their delays and to try again to hit the gap with their Barker code transmissions. Then, the process of block  188  is performed wherein each RU trying to synchronize increments its delay vector and retransmits the same Barker code as was received from the CU. Thereafter, the process of block  185  is performed again wherein the CU monitors the gap for activity. The loop comprising blocks  185 ,  186  and  188 , taken together, comprise the trial and error process which causes all RUs trying to align themselves to continually increment their delay vectors until at least one of them hits the gap. 
     Returning to the consideration of block  197 , if 50% activity level is detected during the authentication interval, it means that only one RU is in the gap. In such a case, the process of block  199  is performed. In this process, the CU identifies the RU whose Barker code transmissions are found in the gap from the unique signature sequence transmitted during the authentication interval. In other words, the CU examines exactly which gaps had correlation peaks therein and the sequence of these gaps and looks up this sequence in a lookup table listing the unique signature sequence for each RU in order to identify the particular RU that has successfully aligned itself. Block  199  is reached only if activity is detected in exactly 50% of the gaps. 
     After the CU identifies the RU, it broadcasts the identity so determined to all RUs as the last step of block  199 . 
     Next, the process of block  200  is performed. In this process, the RU with the identity broadcast by the CU recognizes its identity in the broadcast message and enters a fine tuning mode. The purpose of the fine tuning mode is to cause the value of T d  to be precisely adjusted so that the frame boundaries of frames transmitted by this RU arrive at the CU exactly aligned in time with the frame boundaries of the CU receive frames (which are offset in time from the CU transmit frame boundaries in some embodiments). 
     The fine tuning mode is represented by the process of block  202 . In this process, the CU instructs the RU which has aligned itself in the gap on how to adjust its delay vector in order to center the correlation peak calculated by the CU to the exact middle of the gap. In the preferred embodiment, the gap is comprised of 16 chips which comprise 8 chips in the middle of the gap and then 4 chips on either side of this middle group of 8. It is desirable during the fine tuning mode to get the correlation peak centered in the middle of the middle 8 chips. As mentioned above, a chip is a small interval of time equal to the frame period of 125 microseconds divided by the 448 chips which comprise each frame. In other words, each chip is 279 nanoseconds in duration. The fine tuning process of block  202  involves sending messages back and forth between the CU and the RU which has been identified as having aligned itself in the gap. These messages are sent over the management and control channels. Since clock recovery and carrier recovery has already been accomplished in the RUs before ranging is started, receiving of these management and control messages is no problem and constellations involving phase information can be used. In some embodiments, the exchange involves only one instruction from the CU to the RU saying, for example, “Increase your delay vector by 2 chips” or, “Decrease your delay vector by 3 chips”. In other embodiments, multiple trial and error adjustments are made. The RU then makes the instructed adjustment and retransmits the Barker code. The CU again calculates a correlation peak and examines where the peak occurs in the gap. If the peak occurs in a suitable position, the CU sends a message to the RU telling it to stop adjusting its delay vector as satisfactory alignment has been achieved. The RU then adjusts the coefficients of its precode equalization filters  563  in FIG. 33 to compensate for the phase change caused by the time alignment shift of the fine tuning process. This is done by multiplying all four feed forward coefficients by the negative of the phase shift caused by the timing offset. 
     Returning to the consideration of the process of block  197 , if the CU determines that greater than 50% of the gaps during the authentication interval had correlation peaks therein, i.e., greater than 50% activity is detected, then the process of block  204  is reached. This process is only reached if more than one RU has aligned itself to the same gap. If this case, because each RU is transmitting its unique signature, and because each signature is a unique sequence with only 50% activity level, the result of two RU&#39;s being in the same gap will be that during more than 50% of the gaps of the authentication interval, correlation peaks will occur. It is impossible to find tune the RUs if more than one RU is trying to fine tune during the same gap. Therefore, the CU has to reduce the number of RUs that are in the gap to one, and it starts this process by performing the process of block  204 . In this process, the CU broadcasts a message to all RUs instructing only the RUs attempting to synchronize to execute their collision resolution protocols. 
     Next, the process of block  206  is performed, to start the collision resolution protocol, wherein each RU attempting to synchronize executes a random decision whether to continue attempting to synchronize or to stop attempting to synchronize. Each RU will make this decision with a 50% probability of either outcome. 
     After all RUs make their random decisions whether to continue, the process of block  208  is performed. In this process, the RUs that have decided to continue to align retransmit their signature sequences without changing their timing, i.e., with the same timing as was used on the last iteration of the trial and error process. In other words, each RU that has decided to continue transmits its unique signature sequence (sometimes hereafter called a “dotted sequence”) over another authentication interval using the same delay vectors that are currently set. 
     Next, the process of block  210  on FIG. 5C is performed wherein the CU again monitors the gaps of the authentication interval for activity. 
     If the random decisions whether to continue or not result in no RUs transmitting their signatures, then no activity will be found in the gaps of the authentication interval. In this event, the process of block  212  will be performed wherein the CU broadcasts a message instructing all RUs to go back to the previous stage and to reexecute their decisions to continue or discontinue the ranging process. 
     The RUs then re-execute their decisions whether to continue or stop attempting to align themselves and retransmit their signatures during the authentication interval with the same delay timing used on the previous iteration, as symbolized by block  214 . 
     Following the process of block  214 , the process of block  216  is performed to determine if more than 10 attempts to get one RU in the gap have occurred. If so, the process of block  218  is performed to return to block  181  and restart the ranging process art from the top. If fewer than 10 attempts have been made, processing returns to the process of block  210  wherein the CU again monitors the gaps of the authentication interval for activity. 
     If the process of block  210  finds only one RU in the gap, i.e., 50% activity level is detected during the authentication interval, then the process of block  222  is performed. The process of block  222  authenticates the RU by broadcasting the identity of the RU found in the gap and then the RU is fine tuned in the manner previously described with reference to block  202 . 
     If the CU finds in the process of block  210  more than one RU is still in the gap, processing returns to block  204  where the CU broadcasts a message to all RUs instructing them to execute their collision resolutions protocols. This process is symbolized by block  220 . 
     Alternatives to Preferred Ranging Process 
     There are several alternative embodiments to the ranging process described in FIGS. 5A-5C. They generally fall into two classes. The first class of embodiments is the preferred embodiment represented by FIGS. 5A-5C all of which involve the RU measuring propagation time of its signals to the CU by the trial and error process of adjusting its transmit frame timing delay T d  until a verification management and control message is received from the CU saying “you hit the gap”. There are alternative species within this class wherein the CU sends some kind of an easily detectable marker which triggers the RUs to send some kind of an easily detectable echo signals with good strong correlation peak qualities back to the CU and carrying out the trial and error process to adjust the timing of the echo signals until only one RU is in the gap and a verification message is received from the CU to that effect. In other words, instead of the RU echoing back the same Barker code that the CU sent, the RU could send a chirp or a long, low power sequence that extends over multiple gaps, over an entire frame or over multiple frames. The RU could also send back a very narrow, e.g., one chip wide, high power pulse which is easily detectable over the upstream noise. The CU receiver, during ranging, would perform a correlation on the known chirp, long, low-power sequence, or short, high power pulse to develop correlation peaks. Multiple correlation peaks detected by the CU indicate a contention, and the CU would instruct all RUs that were ranging to “flip the coin” and try again. Once only one RU was ranging and had hit the gap, the identification process would proceed by sending a sequence of whatever signal was sent for initial ranging (or some other easily detectable signal with strong correlation peak characteristics) in a predetermined unique sequence of sequential gaps as in FIGS. 5A-5C. Another alternative species is to perform the trial and error ranging process but eliminating the need for the identification sequence by sending ranging signals which are both easily detectable and unique to each RU. This complicates the CU receiver gap monitor circuit however since it must perform as many different correlations as there are different RUs. This can be done in parallel with a single correlator for each RU or in serial with a single fast correlator that performs multiple correlations on a buffer of samples of the signals received during each (or over whatever is the length of the sequence sent by the RU). Contention would be detected as multiple correlation peaks. Contention resolution would be by a message from the CU to the RUs to flip the coin. Once a single RU was ranging, it would adjust its transmit frame timing delay until it received a message from the CU that its correlation peak had a relative timing relationship to the start of the CU frames such that if the RU transmitter were to transmit with that transmit frame timing delay, its frames would arrive at the CU coincident with the CU frames and all frames of corresponding number from other RUs that were already in frame synchronization. 
     Another alternative embodiment within the class where the RU determines the proper transmit frame timing delay by trial and error generally comprises the following steps. The RU precomputes an 8 of 16 temporary RU ID which is randomly selected. The CU solicits for ranging transmissions. Each RU which wishes to range, transmits its temporary RU ID as 8 Barker code transmissions in 8 gaps of the next 16 RU frames (selected to match the temporary RU ID sequence) with a first iteration of transmit frame timing delay value. The CU generates a ranging status data comprised of 16 bytes, each bit of each byte representing whether a correlation peak occurred during a corresponding chip of the middle 8 chips of a corresponding gap. The CU reorders the 16 bytes into eight 16 bit fields, and transmits this data to all RUs over 4 consecutive frames as a ranging status message which includes data regarding which superframe the ranging status data applies to and the superframe during which the next ranging transmissions are to be made. Each RU receives the status message and stores it in memory and informs the RU computer of the presence of the message. The RU computer parses and scans the ranging status message and interprets the data therein according to the ranging protocol as follows. If all entries are zero, then all ranging RU conclude they have missed the gap and set a new value for their transmit frame timing delays and retransmit their temporary IDs in the next iteration of 16 frames at an activation time specified in the downstream ranging status message. The new transmissions arrive at the CU, and one byte of raw ranging status data is stored in a FIFO memory in the CU. The CU controller initiates a DMA transfer of the FIFO data, and processes the raw ranging data into a new ranging status message and submits valid RU IDs to a training input queue. If the ranging status message analyzed by the RU controller indicates more than one pulse in some gaps, a collision has occurred. If an RU does not find its temporary ID in the status message, it assumes it was involved in the collision, and performs its contention resolution algorithm as described elsewhere herein. If an RU finds its temporary ID in the ranging status data, it is authenticated and in the gap. By looking at the positions of the pulses of its temporary ID in the gap, the RU determines how far off center it is from the middle of the middle 8 chips, and calculates its own offset and applies it to its transmit frame timing delay. The RU is now ready for equalization training. A variation of the above protocol is demand ranging where, after a power failure that would result in all RUs attempting to recover simultaneously thereby swamping the contention resolution mechanism, each RU is addressed individually by its RU ID and asked to begin ranging. 
     The other class of ranging embodiments involves the CU calculating the total turnaround time to each RU and instructing each RU as to how much transmit frame timing delay to use. In this class, the CU sends a marker signal which can be easily detectable by the RU receivers. Each RU trying to range, then immediately transmits back the same easily identifiable signal which can be detected by the CU receiver even if it arrives during the middle of a frame of payload data. Such a signal can be a chirp, a high-power, narrow pulse or a long sequence of chips that spreads out over one or more frames. The CU detects the correlation peak of the signal and compares it to the time of transmission of the original marker signal. The difference is the total turnaround time or TTA. The CU then sends a message to the RU to identify itself which can be done by the “Morse code” authentication a sequence, or in one of the other ways identified above for the first class of ranging embodiments. Once the CU knows the RU&#39;s identity and its TTA, the CU can send a message to the RU instructing it as to how much transmit frame timing delay to use to achieve frame synchronization, and the RU sets this amount of delay for transmission of every frame. 
     Note that in these alternative embodiments of both classes where the ranging signal transmitted by the RU can be detected over the noise of payload data where it arrives at the CU during a frame such as in the embodiments using a large-amplitude, easily detectable pulse or a long sequence which stretches out over one or more frames and which can be detected by a correlator, there is no need for a gap in every frame. The only requirement in high, throughput SCDMA systems is that the RU frames arrive synchronously with correspondingly numbered frames from other RUs (lower throughput CDMA systems do not require frame synchronization). If that timing relationship can be achieved without a gap, then there is no need for a gap. For example, in the case of a narrow, large amplitude pulse, when the RU transmit frame timing delay is set so that this pulse arrives at the beginning of the correspondingly numbered frames from other RUs, then the RU has achieved frame synchronization. In the case of a long sequence that spreads out over, for example, two frames, where the correlation peak is found at the end of the second CU frame, this would mean that if the RU starts a frame transmission at the time it started transmission of the long sequence, that frame will arrive coincident with the CU frame boundaries and therefore, will also be coincident with the frame boundaries of other correspondingly numbered RU frames. Any methodology to achieve this frame synchronization is within the teachings of the invention. 
     Resynchronization When the CU Changes Its Delay Vector 
     The process of adjusting the delay vector used by the CU in transmitting its trigger signal Barker code can result in loss of synchronization by all RUs in the system unless something is done to prevent this before the CU changes its delay. That is, when the CU shortens its delay vector, the RUs closer to the CU than the furthest RU will all go out of alignment unless certain measures are taken to forewarn them of the coming change. There are 3 different embodiments of processes for realigning all of the RUs when the CU changes its delay vector. The preferred one of these embodiments is symbolized by the flow chart of FIG.  7  and involves activity prior to the CU changing its delay vector to prevent loss of synchronization by all RUs when the CU changes its delay. 
     The first of these processes is shown in the flow chart of FIG.  6 . This process will be called the dead reckoning resychronization process for lack of a better term. In this process, the CU concludes, in block  240 , that its delay vector needs to be altered in order to keep the farthest RUs in alignment. This conclusion can be drawn in any one of a number of different ways such as by monitoring the farthest RU for continued alignment after the farthest RU tells the CU that it is aligned with the shortest possible delay vector in use. Or, alternatively, the CU can send out a message to the farthest RU periodically inquiring as to whether it is still aligned. This message can take the form of a request for that RU to transmit its authentication signature and then monitoring the next few frames of an authentication interval to determine if that farthest RUs authentication signature shows up in the authentication interval gaps. If the CU concludes in block  240  that it needs to alter its delay vector it then alters the delay vector. 
     As noted previously, because the CU uses the same delay vector during every frame in transmitting its Barker code trigger signal, the RUs have a predictable periodic signal from the CU upon which they can rely to measure the timing change made by the CU. In other words, the time of arrival of the Barker code from the CU during each frame is predictable to each RU, and when it changes, the RUs can measure by how much it changed. When the Barker code from the CU does not arrive at the predicted time, the RUs know that the CU has just altered its delay vector. The RUs then measure the deviation of the new receive frame timing reference, i.e., the time of arrival of the Barker code trigger signal from the CU, by measuring the difference between the old receive frame timing reference and the new receive frame timing reference. This process is symbolized by block  242 . 
     Finally, each RU realigns itself in the process of block  244 . In this process, each RU alters its delay vector by an amount equal to the change in the receive frame timing reference. Then each RU initiates a ranging process. The CU monitors the gap at the end of every frame so any RU can initiate ranging at any time. 
     FIG. 7 represents the preferred process for resychronizing all RUs after the CU has changed its delay vector. This process will be called the precursor embodiment herein. This process starts with block  246  wherein the CU concludes that it must alter its delay vector to allow the farthest RUs to synchronize to the same frame as the nearest RUs. The a CU, after reaching the conclusion that a change in its delay vector must be made, broadcasts a message to all RUs indicating when and by how much it will alter its delay vector. 
     Next the process of block  248  is performed wherein each RU receives the broadcast and alters its delay vector by an amount equal to the amount that the CU will be changing its delay vector at the specified time. That is, each RU alters its delay vector by the amount instructed by the CU at the time indicated in the message from the CU that the CU will alter its delay vector. 
     Finally, the process of block  250  is performed wherein each RU reinitiates a synchronization process. 
     Both of the embodiments of FIGS. 6 and 7 will result in little or no loss of data because each RU resychronizes very rapidly. This result follows because each RU&#39;s delay vector is immediately set at the delay needed for synchronization at the time the CU alters its delay vector thereby eliminating the delay of the trial and error incrementation of the delay vectors. 
     The final embodiment for resychronizing after the CU changes its delay vector is for the CU simply to broadcast the message to all RUs saying, “You must all now realign as I have just changed my delay vector.” Each RU then re-enters the alignment process symbolized by FIGS. 5A,  5 B, and  5 C. This process is repeated by each RU until all RUs are aligned. 
     Note that in the ranging process described above, it is the RUs that determine how far they are from the CU rather than the CU determining how far each RU is from it. The advantage of having the RUs doing the ranging is that the CU does not have to stop payload traffic on the various channels to perform ranging functions each time a new RU enters the system or an existing RU loses synchronization. In a system where the traffic may frequently include high demand applications such as real time video, stopping traffic flow for ranging is not a viable possibility because it would interrupt the flow of video information and disrupt the subscriber&#39;s video conference, movie etc. In the ranging system described herein in its various embodiments, there is no need to stop traffic since the ranging process is done out of band, i.e., in the gaps. Further, because the transmitted power of the Barker codes is low and correlation processes are used, the process can start blind with any trial and error timing value without interfering with channel traffic. That is, even if the Barker code transmitted back toward the CU by the RU has improper timing and lands somewhere outside the gap, its power level is low enough to not cause substantial interference, and even if some small amount of interference is caused, the chips of the symbols transmitted during the frame have enough redundancy with the Trellis encoded modulation to recover from the interference without an error. Because correlation to a known Barker code pattern (the same Barker code pattern the CU transmitted to the RUs during the previous gap) is used by the CU to determine whether it has or has not detected a Barker code from an RU in the gap, the RUs can transmit their Barker codes at very low power levels so as to avoid interfering with traffic and causing errors in the data of the various payload channels during the trial and error process of setting their transmit frame timing delay values T d  so as to hit the gap. 
     Of course for embodiments where the ranging signals can be detected even when they arrive in the middle of the frame and do not interfere with payload data reception, traffic does not have to be stopped during ranging. As is apparent from the foregoing discussion, there is no need to preset an approximation of the correct transmit frame timing delay into the RUs before they start and then fine tune the delay since even a gross misalignment will not cause any appreciable errors in the payload data. Since Trellis coded modulation and a redundant bit are used in each tribit of payload data, any errors caused by misalignment can be detected and corrected by forward error correction without the need for retransmission. In other embodiments however, conventional ranging techniques could be used where the CU measures the range to the RUs to establish synchronous CDMA, and the particular ranging species initiated by the RUs described herein are not required to practice the invention of a system with a CU with no tracking loops. 
     In the high power pulse embodiments described above, the RUs act like transponders by sending a narrow, high amplitude pulse upon receipt of a trigger signal from the CU. The trigger signal from the CU could be a special pulse, a Barker code, etc. If the RU was misaligned, and the large amplitude pulse landed in the middle of the upstream payload data, the CU would ignore the particular chip which was “stepped on” by the high amplitude pulse. The payload data could still be recovered because the bandwidth of the payload data has been spread so widely using direct sequence CDMA spreading. Trellis code modulation is not needed for this scheme to work. After detecting the RU&#39;s pulse and comparing its timing with the position of the frame timing reference, the CU would ask the RU for its identity and the RU would send it by any conventional manner such as pulse position modulation, amplitude shift keying etc. The CU would then send a message to the RU instructing it to change its transmit frame timing delay in a direction to put the pulse closer to the fixed timing reference, and this process would continue until the RU hit the timing reference. Note in this method, that a gap or guardband is not needed in each frame. 
     Boundless Ranging 
     Note that in the ranging embodiments described above, it is assumed that the “span” of the system, i.e., the difference between the TTA of the farthest RU and the TTA of the nearest RU, is smaller than one frame time. When this is true, all RUs can align to the same gap. When all the RUs are aligned to the same gap, and the CU knows the total turnaround time, dynamic code assignment can be used where the CU informs the RUs by downstream management and control messages what codes each is supposed to use. The CU will then know what codes to use and when to use them in decoding signals from each RU because both the RUs and the CU count frame numbers for the CU frames and all code assignments to the RUs are in terms of CU frame numbers. 
     In very large systems, the span may exceed the frame time, and to force the span to be less than the frame interval would unreasonably constrain the system size. When the span of the system is greater than the frame time, an accounting problem arises because not all the RUs can align to the same gap. This means that the CU will not know which codes each RU used to spread the spectrum of its payload data, unless it knows the total turnaround time to each RU. In other words, each frame transmitted by the CU downstream to the RUs is numbered by virtue of a kiloframe marker signal encoded in the pilot channel carrier tone. The RU receivers detect this kiloframe marker and count individual received frames and thus know what frame number each received frame from the CU is. If the span of the system is less than one frame interval and each RU is aligned to the same gap, each RU will know that when, for example, CU frame  99  is received, the next set of frames transmitted by the RUs all will arrive at the CU at the same time, i.e., the beginning of the next frame at the CU and all those RU frames will have frame number  100  assigned to them by the CU and will be despread and decoded together. In this situation, downstream instructions to RU # 1927  to use codes # 55  and  57  during frame  100  and to RU # 3  to use code # 3  during frame  100  make sense, and the CU can properly decode the data from each of these RUs because it knows which codes each used during frame  100 . Suppose however that RU # 1927  is aligned to the next sequential gap following the gap to which RU # 3  is aligned. This means that when frame # 99  is received from the CU, the frame transmitted by RU # 3  in response to frame # 99  (the downstream data of frame  99  if offloaded, new upstream data is loaded, and the frame is “retransmitted” back toward the CU) will be numbered  100  when it arrives at the CU. However, the frame transmitted by RU # 1927  in response to receipt of frame # 99  will arrive at the beginning of CU frame # 101  and will be treated by the CU as RU frame # 101 . If the CU does not know that RU # 1927  is not aligned to the same gap as RU # 3 , it will assume that RU # 1927  and RU # 3  are both using the codes assigned to them for frame  100 , when RU # 1927  is actually using the codes assigned to it for frame # 101 . 
     One remedy for this accounting problem is for the CU to know the TTA or total turnaround propagation time for each RU and transmit that TTA for each particular RU to that RU. Each RU then uses its TTA time plus the kiloframe marker encoded in the pilot channel (or transmitted downstream in any other way) to keep track of what frame number each received CU frame is and what frame number will be assigned by the CU to the next RU frame transmitted in response to receipt of the CU frame. This allows the RU to use the proper assigned orthogonal, pseudorandom spreading codes assigned by the CU for each frame since the RU will know what frame number will be assigned by the CU to each of the RU&#39;s frames and knows that the code assignment messages from the CU are based upon the frame numbers assigned to RU frames by the CU. 
     The actual algorithm carried out in the CU to calculate TTA for each RU to support boundless ranging in this particular embodiment is quite simple. This algorithm happens after the RU whose TTA is being calculated has successfully completed the ranging process and is aligned with some gap. The CU sends a frame to the RU. The frame itself has no frame tag number, but the kiloframe markers in the pilot channel data allow the RU&#39;s to count received CU frames using a local counter. Meanwhile, as the CU sends frames, its frame count continues to rise. In response to the received frame, the RU sends a frame back to the CU along with a TTA_service_request which includes the RU frame tag number for the transmitted frame which is equal to the local counter value. In other words, the RU frame tag number sent back with the TTA_service request matches the CU frame number of the frame just received as determined by the local counter value. When the RU&#39;s frame reaches the CU, the CU subtracts the RU frame tag number from the CU&#39;s current frame tag count. This difference times the frame interval is equal to the TTA for that RU. The multiplication times the frame interval is not actually necessary since the RU only needs to know how many complete frames behind the current CU frame count each one of the RU&#39;s transmitted frames will be in order to use the proper codes for each frame. 
     High Level Transceiver Block Diagram 
     Referring to FIG. 8, there is shown a high level block diagram of the preferred species of a transceiver for use in the modem of each RU. The CU modem is similar except that it does not have tracking loop circuitry that tracks the carrier and clock signals transmitted by each RU. Instead, the CU circuitry includes circuitry such as that illustrated at  32  in FIG. 1 to periodically correct the phase difference between the master carrier and master clock signals and the carrier signals and clock signals transmitted by each RU based upon preamble data. A block diagram of the CU transceiver is given in FIG.  28 . 
     The circuitry of FIG. 8 that is common to both CU and RU versions will be described below, and the differences between CU and RU versions will be individually discussed where appropriate. If no specific mention is made regarding whether a circuit is in the RU or CU versions, the reader should assume it is identical for both versions. 
     The transmitter  401  of the transceiver uses a framer circuit  400 . The function of the framer is to receive one or more streams of digital data via data path  399  from one or more sources and to organize this data into a plurality of frames, each frame comprised of one or more symbols. In the preferred embodiment, the framer circuit  400  composes the frames of data from a TDMA data stream on bus  399  where each timeslot corresponds to one channel, a channel being a logical construct indicating data on this channel will be kept separated from data on the other channels even though a shared transmission media is used and will appear to the user to have arrived on its own media dedicated solely to it. There are 128 payload data channels to share among all users and there are 16 management and control channels some of which are also shared for a total of 144 channels or timeslots. Each RU may be assigned one or more channels or timeslots depending upon the amount of bandwidth it has been awarded by the CU in response to requests for bandwidth from the RU. In addition, bandwidth may be reserved to the various RUs on a permanent basis in some embodiments, and in these embodiments, the channels or timeslots may be permanently assigned or the reserved number of channels may be assigned on a guaranteed basis each time the RU requests bandwidth. 
     It is not critical to the invention that the incoming data streams arrive in a TDMA stream on bus  399 . The streams of data from peripheral devices or networks could, in alternative embodiments, arrive via FDMA on bus  399  or each source of data could be connected to the framer circuit  400  by a separate input bus. 
     The framer circuit  400  and its associated circuitry implement the variable delay that sets the variable transmit frame timing reference for each RU to achieve frame synchronization by the ranging process described above. This transmit frame timing reference establishes the timing of transmission of the orthogonally CDMA encoded chips of each frame such that all frames arrive from each of the physically distributed RUs at the CU at the same time and aligned with the CU frame boundaries. Although, the invention still works even if frame synchronization is not maintained because of the orthogonality of the CDMA codes which are used, it does not work as well since the maximum number of users which can be simultaneously sharing the available payload channels is limited. This is because there are higher levels of crosstalk between CDMA codes when frame timing synchronization between all RUs and the CU is not maintained. Therefore, each RU undergoes the ranging process described above after first powerup and from time to time thereafter to properly set its transmit frame timing delay to achieve frame synchronization. The transmit frame timing delay T d  is learned by cooperation between the transmitter  401 , the receiver  403  and the CPU  405  in the RU which is ranging and the counterpart devices in the CU by virtue of communication between the RU and CU on the management and control channels. The CPU changes the value of T d  on line  499  until frame synchronization is achieved and thereafter maintains that value of T d  until ranging is performed again. 
     The particular manner in which frame synchronization is achieved is not critical to the invention, and other processes can be used such as by trial and error correlation of a selected data string spread by a selected CDMA code transmitted at differing values of T d  with the signal from another RU which is known to be in frame synchronization which comprises the same data string spread by the same selected CDMA code. 
     The framer circuit  400 , in the preferred TDMA input bus embodiment, bridges the two time domains between the TDMA input data and the chip clock code domain (reading of the framer circuit is done at the chip clock rate and writing is done at the byte clock rate at which timeslots of data are written one 9-bit byte at a time). The output data stream from the framer circuit  400  comprises three arrays of tribits per frame, each array of tribits representing an information vector which, after encoding by the orthogonal multiplexer  408 , is transformed into one symbol of chips. In the preferred embodiment, the orthogonal multiplexer  408  is a code division multiplexer which uses a plurality of orthogonal cyclic codes, each code being used to encode the data from a different channel. This is a so-called direct sequence type spread spectrum operation wherein the bandwidth of the baseband signals on buses  1068 C and  1070 C are spread across a broad spectrum by the CDMA codes using orthogonal code multiplexer  408  in FIG.  8 . This is because of the much higher clock rate of the chip clock used to drive the multiplication of the individual information vector elements times the code elements. 
     In an important class of alternative embodiments, the orthogonal encoding multiplexer  408  could be any encoder which encodes each channel with a different orthogonal waveform for purpose of multiplexing to keep data from different sources separate. For example, these orthogonal multiplexer could store digital samples that define a plurality of orthogonal sine and cosine waveforms, each at a different frequency. Any other set of orthogonal waveforms of different frequencies other than sines and cosines would also work to encode the various channel data samples. Each channel&#39;s data would then be multiplied by a different waveform&#39;s samples to generate new digital samples which define orthogonally encoded data on buses  417  and  419  for modulation onto the RF carrier frequencies. In such embodiments, the bandwidth of each channel&#39;s data is not spread as wide as in a CDMA system. In fact, each channel&#39;s data would be dumped into a narrow bandwidth frequency bin. In such systems, the orthogonal demultiplexer, i.e., decoder  462  in FIG. 8 would perform the inverse transformation on the received samples to bring them back to baseband signals on bus  463 . For example, the orthogonal code multiplexer  408  could be an inverse Fourier transform processor. The inputs to the inverse Fourier transform processor  408  in this alternative embodiment would be the information vector elements on buses  1068 C and  1070 C. Each of these information vector elements would define the magnitude of one frequency component in the Fourier spectrum of the output signal to be generated. The inverse Fourier transform processor would then calculate the time domain waveform that would have that Fourier spectrum and output digital samples that define that time domain waveform on buses  417  and  419 . These samples would be used to modulate one or more RF carriers in accordance with whatever modulation scheme was being used. The receiver&#39;s demultiplexer/decoder  462  on the receiving end of the transmitted signal then performs a Fourier transform on the incoming signal samples to output the individual frequency components the magnitudes of which define the original information vector components. 
     Note that each information vector element in this embodiment always defines the magnitude of the same frequency component. In an alternative multitone system, the information vector elements can be pseudorandomly scrambled in the transmitters so that they define different frequency component magnitudes in each frame and then pseudorandomly descrambled in the same order in the receivers. 
     In SCDMA direct sequence spread spectrum transmitters of the preferred embodiment, the three information vectors output during each frame are converted by CDMA spreading to the three symbols that are transmitted during that frame. The data in each information vector spans the entire 144 timeslots in the sense that three bits from each timeslot or channel are present as the elements of the information vector as a tribit. This interleaving of data from each timeslot into each information vector is preferred but not critical to the invention. Likewise, the transmission of three symbols per frame is not critical to the invention and fewer or greater numbers of symbols could be transmitted. 
     In the preferred embodiment, the circuitry of the transceiver is virtually all digital, so the arrays of tribits are true arrays, the elements of which are used sequentially in the matrix multiplication to perform the CDMA spreading. 
     In analog embodiments, the arrays of tribits will be streams of tribits, with three separate streams per frame. 
     Before finishing the description of the rest of the transceiver circuitry in FIG. 8, the framer circuit  400  will be described in more detail. The RU&#39;s and CU all utilize framer circuitry to implement the delays needed to achieve frame synchronization. The framer is comprised of a FIFO memory and supporting circuitry that stores incoming digital data from the time division multiplexed data stream received by each RU and CU. The difference between the RU and CU framer circuitry is that the CU does not change its transmit frame timing delay except in the situation described above of network expansion which raises the need for the CU to change its delay so that the farthest RUs can synchronize to the same gap as the nearer RUs. The symbols of each frame are composed by outputting the data from the FIFO memory in a different way than it was loaded during each frame. The basic idea is to pass the 9 bit groups of bits from each time slot through the analog of a FIFO delay line implemented by a memory so as to simultaneously implement the delay imposed by each RU and CU needed for frame synchronization while providing a convenient way to compose the symbols of each frame from the data in the TDMA data stream. 
     FIG. 9 shows the circuitry that implements the framer in the preferred embodiment, and FIG. 10 shows the timing relationships between the chip clock signal which sets timing in the code domain and the bit and byte clocks which set timing in the time domain. FIG. 10 also shows a number of other signals generated by time base generator  886 . The basic period from which all other signals are generated is the chip clock signal shown on time line T 1  of FIG.  10 . The relationships between the periods of the various signals in FIG. 10 is shown in parentheses at the right edge of each signal. For example, for the bit clock signal shown on time line T 2  of FIG. 10, for every 7 periods of the chip clock signal, there are 16 periods of the bit clock signal. For every 7 periods in the chip clock signal, there are two periods in the byte clock signal shown on time line T 3  in FIG.  10 . Handling of the TDMA input data stream is synchronized to the bit clock and byte clock signals. 
     The chip clock signal on line  348  of FIG. 9 is generated by a time base generator PLL  886  and is synchronized with the TDMA data stream by the action of the PLL in keeping both the chip clock and bit clock signals synchronized with the master clock reference signal. To this end, the received signal including the Barker code sent by the CU which has the master clock signal embedded therein is fed into the tracking loop of FIG. 11 on line  312 . In the CU receiver, line  312  comes from a from a crystal-controlled oscillator  351  (which is preferably temperature compensated). The crystal oscillator  351  is only present in the CU versions of the modem since the local clock signals generated by the RU modem time bases are phase locked to the recovered master clock signal transmitted by the CU (preferably embedded in the Barker code). Thus, the RU framer circuits receive their time base signals from time base circuits like the circuit  886  in FIG. 8 which is kept synchronized to the master clock signal by the frame detector  513 , PLL  1030  and time base  886  in FIG.  8 . The local chip clock signal of the local clock reference signals on bus  311  are synchronized in phase to the recovered master chip clock signal from the CU. 
     A block diagram of the tracking loop  350  in the RU receivers is shown in FIG.  11 . In FIG. 8, the tracking loop in the RU receiver is shown generally as comprising frame detector  513 , the tracking error signal on line  900 , a voltage controlled oscillator  1030  and time base circuit  886 . In FIG. 9, time base generator PLL is supposed to represent all this tracking loop circuitry. 
     The details of the tracking loop are set forth in FIG. 11. A voltage controlled oscillator  353  operating at a frequency of 14.336 MHz sets the basic operating frequency. The output frequency of the VCO on line  357  is multiplied by a factor or four by multiplier  359  to generate a high speed clock signal at 57.344 MHz on line  367 . This oscillator  353  corresponds to VCO  1030  in embodiments like that shown in FIG. 8. A bit clock signal on line  377  is synthesized by dividing the frequency of the high speed clock signal on line  367  by a factor of 7 in a divide-by-seven counter  369  to generate a bit clock signal having a frequency of 8.192 Mhz. A chip clock signal on line  348  is generated by dividing the high speed clock signal on line  367  by a factor of 16 in a divide-by-16 counter  371  to generate a chip clock signal having a frequency of 3.548 Mhz. The multiplier  359 , divider  369  and divider  371  together corresponds to the time base circuit  886  in embodiments like that shown in FIG.  8 . 
     The bit clock and chip clock signals are kept synchronized in RUs to the master clock signal by a phase detector  373  which compares the phase of the received signal on line  312  to the phase of the bit clock signal and outputs a signal which is coupled to the frequency control input  375  of the VCO through a low pass filter  397 . The bit clock and chip clock signals in the CU modem are kept synchronized to the master clock signal on line  312  from a crystal control oscillator. In the RUs, the phase detector  373  takes the form of the clock recovery circuits in the frame detector described elsewhere herein coupled with a phase detection circuit that compares the phase of the recovered clock to the phase of the bit clock signal. The phase detector circuit  373  and low pass filter correspond to the frame detector circuit  513  in the block diagram of the RU transceiver shown in FIG.  8 . 
     The relationships between timing in the time domain and timing in the code domain are as follows: 
     There are 144 total time slots or channels in the TDMA stream, of which 128 are payload time slots and 16 are management and control time slots; 
     Each time slot or channel in the TDMA streams carries 9 bits of digital data synchronized with the bit clock; 
     One time slot worth of data or 9 bits is stored in the framer for each cycle of the byte clock; 
     1 frame=144 times slots, each with 9 bits plus 16 chips for the alignment gap; 
     1 frame also equals 3 symbols plus the 16 chip periods of the alignment gap=448 chip periods; 
     1 symbol=144 chip periods; 
     1 gap=16 chip periods; 
     For every 16 bit clock periods, there are 7 chip clock periods, and for every byte clock period, there are 9 bit clock periods. 
     To implement the delay necessary in each RU and CU transmit channel circuitry to maintain frame synchronization, consider the following with reference to FIG.  9 . The data stream coming into the framer circuitry during each time slot is stored in a different address in memory  300  in FIG. 9 at the data rate of the byte clock signal on line  302 . The byte clock signal on line  302  is generated by a byte counter  370  shown at the top of FIG. 9 which generates a byte clock signal transition on line  302  every 9 cycles of the bit clock signal on line  377  from the time base generator  350 . Memory  300  is a three page memory and the addressing circuitry of FIG. 9 controls the address and data ports such that data is written into and read from the two pages by alternating the use of these buses. Data from the time slots/channels in the time division multiplexed stream of serial data on line  301  is shifted serially into a serial-in, parallel-out shift register  310  at the bit clock rate of the signal on line  377 . The byte clock signal on line  302  causes a register  314  to store the current 9-bit, parallel format output of the shift register on bus  316  after each 9 new bits are shifted into shift register  310 . 
     The 9 bit parallel format output of the register  314  is presented on 9-bit bus  318  to the write data input port of memory  300 . Thus, a new 9-bit group of data from the TDMA stream is presented for storage on each cycle of the byte clock signal. Each 9-bit group of data from the TDMA stream is stored in a different memory location of memory  300  as will become clear from the discussion of the address generation circuitry described below. 
     Data is read out of memory  300  at the same rate at which it was stored, but starting at some programmable time after the data is stored, thereby implementing the variable delay needed to maintain frame synchronization with the CU frame timing. This programmable delay T d  is set by the difference in addresses between the address stored in a receive frame counter (read pointer) and the address stored in a transmit frame counter  324  in FIG. 12 (write pointer). 
     To illustrate this concept, FIG. 13 represents portions of memory  300  with the stippled portions  306  and  308  representing the number of addresses difference between the position of the read pointer and the position of the write pointer to implement the delay T d . The cross hatched portion  304  represents one frame of 9-bit bytes while the stippled portions  306  and  308  represent the amount of the delay T d , where portion  306  represents a portion of the delay T d  expressed in full 9-bit bytes, and portion  308  represents the remainder of the delay T d  expressed as part of a single byte. In other words, the delay T d  may be some fraction of the number of bit clocks making up an entire 9-bit byte. This is because the delay needed to maintain frame synchronization may not work out to be an integer number of byte clocks. 
     FIG. 9 shows how the time delay T d  is implemented using a receive frame counter  322  that generates the write pointer address controlling where incoming data is stored in the memory  300  and a transmit frame counter  324  that generates a read address pointer that controls the read address from which data is read for transmission. The F_sync signal on line  326  resets the write pointer in counter  322  to zero at the beginning of each new frame. A modulo adder  326  adds the number of chip clocks based upon the desired time delay T d  to the output write pointer on bus  328  and inputs the result into the transmit frame counter  324  as the read pointer. The value of T d  is varied on a trial and error basis during the synchronization process until the gap is hit and the CU sends a message to whatever RU is synchronizing telling it to freeze T d  at the value that caused the gap to be hit by the Barker code. 
     FIG. 14 is a memory filling diagram that illustrates how entire 9-bit bytes are received continuously, while 3-bit tribits for each of 144 channels are sent out simultaneously to compose the symbols of each frame. FIG. 14 graphically illustrates how the frame memory  300  fills and is emptied during this process. Frame memory  300  has 144 memory locations corresponding to the 144 channels of the system on each of three pages. While one page is being filled, another page is being simultaneously emptied at the same rate. Each memory address on each page can store the 9 bits of data from one of the 144 time slots in the TDMA stream. 16 memory locations on each page are reserved for the storage of management and control data to be sent across the 16 management and control channels. In FIG. 14, address numbers increase with an increasing Y coordinate. 
     At time ( 0 ) in FIG. 14 (the leftmost column), page one of the memory is shown as completely full with one frame of data comprised of three vertical columns of three cross-hatched blocks apiece. Each column of three blocks, such as blocks  334 ,  336  and  338  represent one symbol, each symbol having 48 tribits therein. The middle column of FIG. 14 represents the state of fill of the memory after transmission of the first symbol comprised of blocks  334 ,  336  and  338 . The rightmost column of FIG. 14 represents the state of fill of the memory after transmission of symbol  2  comprised of blocks encircled by dashed line  334 . 
     The width along the X axis of each individual crosshatched block in FIG. 14 is equal to the 3 bits of a tribit, and the entire width of a column of blocks is equal to the 9 bits of a time slot. The positive x direction represents increasing time in the time domain. In other words, the first 9-bit byte that is stored is stored in the lowest row of the lowest three blocks in the left column with increasing time in the TDMA stream extending from left to right. 
     The blocks surrounded by dashed line  332  in the leftmost column represent 144 memory locations, each storing the 9 bits from one of the 144 time slots in one frame of data. The three crosshatched blocks  334 ,  336  and  338  represent the first symbol of the first frame, each symbol storing 48 tribits. Note in the middle column, after transmission of the first symbol in the frame, these three blocks are gone. Note also that the data of symbol  1  is read out of the memory “across time”, i.e., along the y axis, thereby interleaving the data from the first tribits of individual channels in the time domain into different temporal relationships in the code domain and spreading out the energy of the time slot data over the entire frame interval. This is part of the teaching of code division, multiple access or CDMA modulation schemes, but is not critical to the invention. Interleaving of data improves the immunity of the data to burst noise. 
     The three blocks within dashed box  334  in the middle column of FIG. 14 represent the second symbol of data that is to be transmitted in the first frame. Note that these three blocks are gone in the rightmost column representing the state of page one of the memory fill after transmission of the second symbol. 
     While the first and second symbols are being transmitted, another page of the memory  300  continues to fill up as the data from new timeslots is received. For example, while symbol  1  from page  1  of the memory is being transmitted during the first frame, the data in the three blocks encircled by dashed line  336  in the middle column is received in page  2  of the memory and stored. Thus, while one third of the data from page  1  is read and transmitted, one third of page  2  of the memory is filled with new data. Likewise, while the second symbol of page  1  is being transmitted, the data represented by the three blocks encircled by dashed line  338  in the right column is received and stored in page  2  of the memory. 
     The blocks encircled by dashed box  340  represents the delay T d  implemented by modulo adder  326  in FIG.  15  and the 16 chip alignment gap. 
     FIG. 15 is a diagram of the relative rates of address incrementation of the read and write pointers used to manage the framer buffer memory  300  including the relative timing of address incrementation for reading the tribits. Dashed line  342  represents the rate of address incrementation of the write pointer generated by counter  322  in FIGS. 12 and 15. This counter counts transitions in the byte clock signal on line  302  in FIG. 9, with the byte clock signal shown on time line T 3  in FIG.  10 . Every cycle of the byte clock signal causes register  314  in FIG. 9 to latch a new 9-bit byte therein and present it on bus  318  to the write data port of two-port memory  300 . Every cycle of the byte counter also causes write pointer counter  322  to present a new write pointer address on bus  366  for use in controlling where the data on bus  318  is stored. A multiplexer  362  having its output coupled to the address port of memory  300  and having as its inputs the write pointer on bus  366  and the read pointer on bus  364  is suitably switched so that the write pointer and read pointer addresses are presented at the appropriate times at the address port to implement the memory filling and memory reading operations described herein. 
     The bit clock signal on line  377  in FIG. 9 is used to clock the serial-in, parallel out shift register  310 . The bit clock signal is generated by the time base generator shown in FIG.  11  and is counted by a modulo 9 bit counter  372  shown at the top of FIG. 9 for purposes of helping generate the byte clock signal on line  302  in FIG.  9 . This counter  372  counts the bit clock signal on line  377  from time base generator  866  modulo 9 and outputs a transition to logic 0 on line  374  after every 9th bit period. The transition on line  374  acts as a count enable signal to byte counter  370  to enable incrementation of the byte counter  370  by the next bit clock cycle. This generates the byte counter signal on line  302 . The bit counter  372  is always enabled by the hard wired count enable signal on line  376 . Both the bit counter and the byte counter are reset to 0 by asserting the F 0  signal on line  299  for fast resetting and resynchronization of the system. The F 0  signal occurs at the end of each frame. The F 0  signal is generated by a portion of the time base generator not shown in FIG. 11, and is counted as a clock signal by frame counter  376  which outputs a synchronized F 0  signal on line  299 ′. The frame counter  376  is reset every 4th frame by a super frame signal F 4 . 
     The time delay T d  necessary for hitting the CU gap with a Barker code transmission is added to the F 0  signal on line  299 ′ by the modulo adder  326  to generate the F 0 ′ signal on line  381 . The value of T d  is received from the CPU via bus  499  and changes by trial and error during ranging but is frozen at whatever delay centers the transmitter Barker code in the gap at the CU. The F 0  signal on line  299 ′ also increments the page pointer  321  for the write pointer and simultaneously resets the write pointer  322  to zero at the end of each frame so as to cause a page swap and begin writing again at address 0 of the next page. 
     The delayed F 0 ′ signal on line  381  increments the page pointer  323  of the read address circuitry to cause a page swap and simultaneously resets the read pointer counter  324  to zero so as to begin reading at address 0 of the next page at the end of the frame. 
     Returning to the consideration of FIG. 15, solid line  346  represents the rate of emptying the frame memory  300  in FIG.  9 . This rate of emptying is based upon incrementation of the read pointer counter which counts the chip clock signal on line  348  from time base generator  866 . Since each symbol stores 144 tribits from 144 different channels and since there are three symbols and a 16 chip gap in each frame, the total number of chips in a frame is  448 . Since all the 432 tribits of all three symbols of the frame must be read out while the byte counter is counting to 144 to store a frame&#39;s worth of 9-bit bytes of data from 144 channels or time slots, the read pointer is incremented on the chip clock signal. This causes all 432 tribits from all three symbols of a frame to be read out while the next frame of data is being stored thereby preventing overflow of memory  300 . This is why the read pointer line  346  in FIG. 15 is shown as emptying the memory at the same rate as the write pointer fills it. 
     Line  352  in FIG. 15 represents the rate of incrementation of the read pointer counter  324  in FIG.  9 . The read pointer counter increments on each cycle of the chip clock signal such that it increments from 0 to 143 during the time to read all the tribits from the first symbol. This has the effect of causing the 9 bits of data from each of the 144 timeslots or channels to appear sequentially at the read data output bus  358 . However, it is desired to only unload all 144 tribits from a single symbol during one symbol time, so some switching on the output bus is needed, as described below. 
     A tribit select counter (which is not shown in FIG. 9) is coupled with a multiplexer  356  which does this switching. This tribit select counter generates a tribit select signal on line  354  in FIG. 9 which controls switching by a multiplexer  356 . This multiplexer has an input coupled to the 9-bit read data output port  358  of the memory  300 . The tribit select counter counts at a rate to generate the select signal on line  354  in such a way as to cause only tribits from the first symbol to be output from the multiplexer  356  on bus  360  during the time that first symbol is being transmitted. 
     FIG. 16 is a diagram which helps illustrate the manner in which framer memory  300  is emptied for transmission. FIG. 16 shows a completely filled page  1  of memory  300  in FIG. 9 comprising 144 memory addresses, each filled with one 9-bit byte, and divided into three columns of 3-bit tribits. Each column, marked by the legends symbol  1 , symbol  2  and symbol  3 , is comprised of 144 tribits and represents one symbol of a frame. To send this frame of data, the read pointer will increment 144 times during the time the first symbol is being encoded. The state of the tribit select counter during this first 144 cycles is such that only the 144 tribits of symbol  1  will be output on bus  360  to the forward error correction (FEC) encoder  402  in FIG.  8 . 
     After the 144th incrementation, the read pointer counter  324  rolls over to zero and begins to count up to 143 again. At the 144th incrementation, the tribit select counter increments which causes the multiplexer  356  to select the middle column of tribits from symbol  2  in FIG. 16 for output on bus  360  in FIG. 9 to the forward error correction encoder  402  in FIG. 8. A similar process unloads the 144 tribits of symbol  3 . 
     Bus  360  in FIG. 9 is, in the embodiment shown in FIG. 8, coupled to a forward error correction encoder  402 . In FIG. 9, a multiplexer  362  having its output coupled to the address input of the framer memory  300  has two inputs: one is coupled to the output of the read pointer counter  324  and the other is coupled to the output of the write pointer counter  322 . This multiplexer alternately couples the read pointer on bus  364  and the write pointer  366  to the address port  368  of the memory  300  on every cycle of the chip clock signal on line  348 . The chip clock signal is also coupled to the control input of the memory  300  to serve as the RD/WR* control signal controlling whether the memory uses the address at port  368  in a read or a write transaction. 
     Returning to the consideration of the transceiver block diagram of FIG. 8, the output data streams from the framer on bus  360  in FIG. 9 may optionally be passed through a forward error correction encoder  402 . The forward error correction encoder  402  can be eliminated in some embodiments or an ARQ encoder may be substituted. The embodiment of FIG. 8 symbolizes a class of species which use systematic codes where the bits of the tribits are not scrambled and the FEC encoder is a convolutional encoder. In alternative embodiments, the tribits on bus  360  can be pseudorandomly scrambled prior to being received by the FEC encoder  402 . In other alternative embodiments, the FEC encoder can use block codes. In the preferred embodiment, FEC encoder  402  is used for Trellis encoding. 
     The purpose of the forward error correction encoder  402  is to add one or more redundant bits to each tribit so as to improve the error rate for the energy per bit-to-noise power density ratio resulting from the chosen modulation scheme. In the preferred embodiment, the FEC encoder  402  is a Trellis encoder for a 16-QAM, Rate 3/4 Trellis code having 16 states, a pi/4 rotational invariant, no parallel paths and an effective code length of 2. In yet another alternative embodiment, the forward error correction encoder  402  could be a Reed-Solomon Encoder which generates a first set of code words which are then further encoded in a Trellis encoder. An advantage of using Trellis encoded modulation either with or without Reed-Solomon coding is that it allows redundancy to be added to the payload data so as to enable forward error correction without increasing the symbol rate and the consumed bandwidth. This prevents the need for retransmission of garbled data since the errors can usually be eliminated by the Viterbi decoder using the redundant bits. Trellis encoded modulation uses redundant bits to map the payload data into a larger constellation of possible points (called signal space coding). The bandwidth required for transmission is not increased, nor is total noise admitted by the receive filter. Basically, Trellis encoding uses a channel coder to receive each k payload bits and convert them into n bits where n is greater than k and includes some redundant bits which contain information about the k payload bits. The n bit group is then processed by a modified line coder to produce symbols for transmission from a constellation having size 2 n . Significant coding gains can be achieved in this way. For example, assuming a particular additive white Gaussian noise channel produces an acceptable probability of error without coding at some signal to noise ratio using a constellation of size M, using Trellis encoded modulation, the error probability can be reduced at the same signal to noise ratio or the signal to noise ratio can be reduced at the same error probability, and, per Ungerboeck, most of this theoretical reduction can be achieved using a constellation of 2M plus a channel coding scheme. As an example of the type of coding gain that can be achieved using Trellis coded modulation, consider the following. If only tribits were used without coding with redundant bits, and an 8-AM constellation were used, according to Ungerboeck transmission with 10 − 5 error probability with an SNR of 26 transmitting and 3 bits per symbol could be done. However, by the use of Trellis encoded modulation using a 16-AM constellation, it is possible to send 3 bits error free down to 18 db SNR. Therefore, using Trellis encoding, it possible to achieve a coding plus shaping gain of 26−18=8 db. In the invention, a coding gain of approximately 4 db is obtained. The main advantage of using Trellis coded modulation is the ability to reduce the error rate or increase the number of payload bits without increasing the symbol rate and bandwidth consumed. This can be done using a constellation no greater than 2M. More details about Trellis encoded modulation are contained in Lee and Messerschmit,  Digital Communication,  2 d Ed.,  1994 (Kluwer Academic Publishers, Boston), ISBN 0 7923 9391 0, which is hereby incorporated by reference. Trellis encoded modulation is not required however to practice the invention of a CU with no tracking loops to constantly track the RU clock and carrier frequencies, and, therefore, the encoder  402  in FIG. 8 could be eliminated or replaced with simple encoders using any known error detection or correction encoding scheme and a mapper to map the resulting encoded symbols into points in a constellation. 
     In the preferred embodiment, the forward error correction encoder  402  take the form of the Trellis encoder shown in FIG.  17 . The input to the encoder is comprised of three payload bits of a tribit on lines W 1 , W 2  and W 3  of bus  509 . Bits W 3  and W 2  pass through the convolutional encoder section unchanged and arrive at mapper  1050  unchanged as bits y 3  and y 2 , respectively. Bit y 0  at the input of the mapper is generated by an encoder section comprised of D flip flops  1052 ,  1054 ,  1056  and  1058  coupled by exclusive-OR gates  1060 ,  1062  and  1064 . These exclusive-OR gates combine the outputs of the flip flops with various combinations of the W 3 , W 2  and W 1  bits and two feedback bits. The output of flip flop  1058  is the y 0  bit and is a factor in the generation of the two feedback bits. The y 1  bit is the W 1  bit after an exclusive-OR operation in a precoder  1066  with the output of flip flop  1056 . 
     Mapper  1050  has a normal mode and several other modes including a fallback mode. In normal mode, the mapper takes the 16 combinations of the y 0  through y 3  bits and maps them to the 16-QAM constellation of FIG.  18 . The mapper outputs 2 bits on an I bus  1068  and 2 bits on a Q bus  1070 . For input combination of 4 bits, the two bits on the I bus define the coordinate along the I axis in FIG. 18 of the resulting constellation point, and the two bits on the Q bus define the coordinate along the Q axis. The mapping is nonlinear, and is defined by the table of FIG.  19 . For example, an input code of  0101  for bits y 0 -y 3 , maps to a 1+3*j constellation point having an I coordinate of 1 and a Q axis coordinate of +3. This Trellis encoder has a code gain of approximately 4 db SNR. 
     The I and Q bits on buses  1068  and  1070  are then stored as separate real and imaginary arrays for the information vector [b] in memory  406  in FIG.  8 . These real and imaginary arrays then have their bandwidths spread individually by CDMA orthogonal code multiplexer  408  in the manner illustrated by FIG. 20B to generate real and imaginary array components of a result vector. The elements of each result vector define the individual chips of one symbol in a frame. 
     Fallback mode and the other available modes are implemented in the Trellis encoder of FIG.  17  through control signals on bus  1072  from CPU  405  in FIG.  8 . The mapper has normal mode, fallback mode, access channel mode, training channel mode and no code mode in some embodiments. In fallback mode, the encoder output in divided into two symbols and transmitted separately. The 2 LSBs (y 0 , y 1 ) are transmitted as the first symbol and the 2 MSBs (y 3 ,y 2 ) are transmitted in the second symbol. The 2 LSBs are transmitted QPSK with a 4 point constellation. The two MSBs are transmitted DQPSK. To avoid changing the output power during fallback mode, the 4 point constellation of FIG. 21 was chosen for fallback mode. FIG. 22 shows the mapping for the LSB and MSB chips in fallback mode. The receiver needs to be synchronized between the first and second symbols to know which symbol carries the information of the LSBs and MSBs. In other words, when the transmitter goes into fallback mode, the 144 tribits that were encoded and then mapped into the 144 chips of the first symbol in the first frame in normal mode are now split into LSB and MSB components where are mapped into the first and second symbols of the first frame in fallback mode. Likewise, the 144 tribits that were encoded and mapped into the second symbol of the first frame in normal mode are split and mapped into the third symbol of the first frame and the first symbol of the second frame. Since the receiver is synchronized and knows which symbol of which frame it is receiving at all times, the CPU  405  controls the deframer circuit  470  so as to properly reassemble the original data stream via signals on bus  1076  in FIG.  8 . Constant link quality monitoring for noise, crosstalk and signal quality is performed in background cycling constantly through all codes and timeslots. When a fallback mode threshold is exceeded, fallback mode is initiated and maintained until conditions return below threshold. Monitored values are stored by a diversity management function in the CU computer that controls code diversity and fallback operations. 
     In embodiments where forward error correction is not used, encoder  402  is an ARQ encoder which simply adds enough ECC bits to allow the receiver to detect an error and request a retransmission. The retransmission request is made on one of the command and control channels. In some block code embodiments, the forward error correction encoder  402  uses cyclic codes where the sum of any two code words is a code word and any cyclic shift of a code word is also a code word. Note that the Viterbi decoder  468  discussed below in the description of the receiver is used only when the forward error correction encoder  402  is a convolutional or Trellis encoder. 
     Although the discussion of the forward error correction encoder  402  has not heretofore included any discussion of the modulation process carried out by modulator  410 , Trellis-Coded Modulation (hereafter referred to as TCM) is preferred because of its lower error rate in the face of channel impairments. TCM modulation combines the forward error correction and modulation process by redefining the coding as the process of imposing certain patterns on the transmitted signal. This provide more effective utilization of band-limited channels as is the case for multiple access on HFC cable TV plants. Trellis-Coded Modulation is characterized by three basic features: 
     (1) the number of signal points in the constellation used is larger than what is required for the modulation format of interest with the same data rate wherein the additional points allow redundancy for forward error control coding without sacrificing bandwidth; 
     (2) convolutional encoding of the message data is used to introduce a certain dependency between successive signal points such that only certain patterns or sequences of signal points are permitted; and 
     (3) soft-decision decoding is performed in the receiver, in which the permissible sequence of signals is modelled as a Trellis code. 
     The preferred form of the encoder  402  is the 16 state Trellis encoder shown in FIG.  17 . This encoder is characterized by parity check polynomials given in octal form as follows: h 3 =04, h 2 =10, h 1 =06, h 0 =23, d{circumflex over ( )} 2 _free=5.0, Nfree=1.68. The nonlinear term is given by D{circumflex over ( )} 2 [y 0 (S).AND.D{circumflex over ( )}(−1)y 0 (D)]. More details are given in Pietrobon, Ungerboeck et al., “Rotationally Invariant Nonlinear Trellis Codes for Two Dimensional Modulation,” IEEE Transactions on Information Theory, Vol. 40, No. 6, November 1994, pp. 1773-1791, which is hereby incorporated by reference. 
     In the preferred embodiment, the forward error correction encoder  402  has multiple modes which add different numbers of redundant bits while always maintaining the code word length at 4 bits. In a normal mode, one redundant bit is added per tribit. In a fallback mode when channel impairments are high, fewer payload bits are sent and more redundant bits are sent in each 4 bit code word. 
     The encoder  402  in the transmitter is a state machine which, in conjunction with state memory  404 , receives the stream of tribits for each symbol and calculates a 4th redundancy bit for each tribit. This 4th bit provides redundancy for error detection and correction and for use by a Viterbi Decoder  468  in the receiver in ascertaining with greater accuracy the data that was actually sent despite the presence of noise. The 4th bit in each tribit is part of the Trellis modulation scheme and is generated by the convolutional encoder  402 . A three bit constellation would normally have only 8 points. However, Trellis modulation adds redundant bits interspersed in the information stream of tribits and increases the size of the constellation to enable more spacing between constellation points thereby enabling better discrimination between points by the receiver and lowering the bit error rate without increasing the bandwidth. In noisy environments like CATV media, Trellis modulation is preferred, but some species of the invention will work without the redundant 4th bits and using a smaller constellation. In the preferred embodiment, the encoder is used to provide greater accuracy and better noise immunity. The encoder, in the preferred embodiment, is a state machine but it could also be a lookup table implemented in RAM or ROM etc. The implementation of the state machine is not critical as long as the implementation is fast enough to keep up with the chip clock data rate. For purposes of this discussion, it will be assumed that the convolutional encoder  402  is present. 
     M-ary Modulation in Code Division Multiple Access System 
     The output of the convolutional encoder  402  is an array of 4-bit digital numbers for each of symbols  1 ,  2  and  3  shown in FIG.  2 A. Each of these 4-bit numbers has two bits representing a real part and two bits representing an imaginary part. Thus, the information vector [b] shown at  481  in FIG. 20A for use in the matrix multiplication for CDMA spreading of each symbol is comprised of 144 4-bit elements, each element comprising one tribit plus the additional 4th bit calculated by the convolutional encoder  402 . Each 4-bit symbol element in FIG. 20A, such as element  483  represents one third of the information bits from the corresponding timeslot in the TDMA stream input received by the transceiver plus the redundant bit calculated by the convolutional encoder  402 . FIG. 20A illustrates how the information vector [b] for each symbol has its energy spread over time by the process of code division multiplexing implemented using matrix multiplication of the information vector [b] of each symbol times a matrix of orthogonal codes. The first two bits of each 4-bit symbol element are used to define the amplitude of either the I or Q coordinate, and the last two bits are used to define the amplitude of the other orthogonal vector element. The constellation of input point mappings of all possible points defined by a 4 bit symbol element or “chip” is shown in FIG.  18 . FIG. 18 maps each of 16 possible input points, i.e., permutations of the 4 bits of each chip in each symbol array to a point in space defined by the in-phase or I axis for the real part and the quadrature or Q axis for the imaginary part of each point. The I coordinate of each point represents the amplitude for that point imposed upon the sine wave carrier fed to the modulator  410  in FIG. 8 on line  427  (only the COS signal is shown and the sine wave is generated internally to the modulator  410  by performing a 90 degree phase shift) to modulate that point. The Q coordinate of each point in the constellation represents the amplitude imposed by modulator  410  on the cosine wave carrier fed to it in order to modulate the point in QAM Trellis modulation. FIG. 19 is a table listing all the possible 16 combinations of 4 bits in the Code column and the corresponding 2&#39;s complement digital representation of the real and imaginary coordinates for each combination in the Inphase and Quadrature columns, respectively. For example, the input point  1100  maps to a point having a +3 imaginary coordinate and a −1 real coordinate on the constellation of FIG.  18 . The mapping of FIG. 18 was selected to give maximum separation between points in the constellation for best noise immunity, but any other mapping would also work. Likewise, 2&#39;s complement representation is not required for the coordinates as they can be represented in other number systems as well. In the preferred embodiment, the encoder  402  is a Trellis encoder coupled to a state memory  404 . The function of the Trellis encoder  402  is to select the bit to append to each tribit to put it at a place in the 16 point constellation of FIG. 18 which gives maximum noise immunity. This selection is made according to known Trellis modulation principles based upon the previous states. In other words, Trellis encoder  402  and state memory  404  comprise a state machine which transitions to one of the 16 states or points in the constellation based during each chip time based upon the incoming tribit data and the previous states. The memory  404 , in the preferred embodiment, is large enough to record the last state for each of the time slots, so as each tribit arrives, the last state for the time slot from which the tribit was generated is looked up in memory  404 , and the tribit is encoded based upon that channel&#39;s prior state. 
     The stream of 4-bit symbol elements that are output from the encoder  402  are stored in memory  406  as three different linear arrays corresponding to symbols  1 ,  2  and  3  in FIG.  16 . Each 4-bit symbol element is a complex number comprised of 2 bits which define the magnitude of the I or inphase coordinate of a constellation point and 2 bits which define the magnitude of the Q or quadrature coordinate of the same constellation point. These two I and Q values are output on buses  1068  and  1070 . 
     After passing the tribit stream from the framer  400  through the encoder, the resulting 4-bit data streams are stored as separate I and Q information vector arrays for each symbol in memory  406 . Each symbol is comprised of two linear arrays of 2 bit numbers: one array contains multiple 2-bit elements defining the real or inphase “I” coordinates for all the elements of the symbol and the other array stores the 2-bit elements which define the imaginary or quadrature “Q” coordinate of each symbol element. The 144 array elements of each symbol define an information vector b for each symbol. The code division multiplexer  408  then spreads each information vector separately with a separate orthogonal code for each channel and combines the spread data into a single orthogonally coded data stream. 
     FIGS. 20A and 20B show the matrix multiplication process which is performed within code division multiplexer  408  in FIG. 8 to multiply each of the two linear arrays that define each symbol times the orthogonal code matrix [c] identified as matrix  407  in FIG.  20 B. In the preferred embodiment, the matrix multiplication is performed by a microprocessor, but any machine that can do the matrix multiplication will suffice to practice the invention. 
     The encoding in CDMA MUX  408  spreads the energy of the symbols over time using orthogonal codes or orthogonal, cyclic codes. This is done in two steps. First, a linear array information vector of just real parts, i.e., inphase coordinates of the symbol to be transmitted, symbolized by array  405  in FIG. 20B, is multiplied by the code matrix  407 . This operation generates another linear array of real or inphase coordinates along the R axis of a result space in a results constellation similar to the constellation of all possible input points shown in FIG.  18 . This first linear array  409  defines the real axis coordinates in the result constellation for a plurality of chips from the first symbol to be transmitted. 
     Second, the same process is repeated for the imaginary coordinate linear array (not shown) for the same symbol the real coordinates of which were just processed. This results in another linear array comprising the imaginary or quadrature coordinates of the chips in the results array. This imaginary component array of the results array also is not shown in FIG.  20 B. 
     The real component array, represented by linear array  409 , is part of an overall result or “chips out” array which contains both the real and imaginary coordinates of a plurality of chips to be transmitted. These chips map to points in the result space, and the points in the result space map to whatever points in the input point space that are defined by the real and imaginary components in the information vector array b, of which array  405  is the real part. The mapping between the input point space and the results space is defined by the contents of the code matrix and the orthogonal codes. 
     Before performing the matrix multiplication, the 2&#39;s complement values of the real and imaginary components of the information vector b input array are converted to their decimal equivalents as shown in FIG. 20B in some embodiments. FIG. 20B is a simplified version of the system in which there are only 4 channels resulting in 4 elements of each symbol. The 4 real components of the information vector b shown in array  405  after conversion to their decimal equivalents, are, respectively from top to bottom, +3 (first three bits of channel  1 ), −1 (first three bits of channel  2 ), −1 (first three bits of channel  3 ) and +3 (first three bits of channel  4 ). This column of numbers is multiplied by the first row in the code matrix to yield the result 4 as the first real component in results array  409 . This result is derived from summing the partial products as follows [(3×1)+(−1×1)+(−1×1)+(3×1)]=4. The next component down in the real part array  409 , i.e., 0, is derived by multiplying the next real component down in the array  405  (−1) times the second row of the code matrix in a similar manner yielding [(−1×1)+(−1×1)+(−1×1)+(−1×1)]=0. In the preferred embodiment, arrays  405  and  409  would be 144 elements long, and the code matrix  407  would have 144 elements in each row and would have 144 rows. The orthogonal codes are actually the columns of the array. Note that the channel  1  element always gets multiplied by an element of the first column and so on for all the elements of array  405  as array  405  is multiplied by each of the 4 rows in array  407 . Thus, the first column in array  407  is the orthogonal code used to spread out the bandwidth of the data from the channel  1  timeslot. For ease of generation, the set of orthogonal pseudorandom codes in matrix  407  is also cyclic. 
     Because each orthogonal code used in array  407  is also pseudorandom, and the rate of generation of the chips in the result vector (the chip rate) is much higher than the bandwidth of the input data represented by the information vector  405 , the bandwidth of the resulting signals defined by the result vectors generated by this process is spread into an extremely broad spectrum. In fact, the bandwidth of the result vectors generated by this process extends to plus and minus infinity. The spread signal consists of replicas of the same power spectrum repeated end to end, so the signal can be recovered by the receiver even though only the portion within the passband of the amplifiers on the hybrid fiber coax channel and the transmitter and receiver filters is processed by the demodulation and despreading circuitry in the receiver. 
     The CDMA MUX  408  in FIG. 8 that does the matrix multiplication can be a programmed microprocessor or a dedicated custom logic circuit, etc. Any design which can perform the multiplication of the information vector times the code elements for all the active channels will suffice. Since the code matrix is comprised of purely 1&#39;s and −1&#39;s, the multiplication is made simpler. If the codes in the code matrix are Hadamard codes, the matrix multiplication can be made using the Fast Hadamard Transform algorithm in a digital signal processor or microprocessor. If the code matrix is comprised of sin and cosine terms, the Fast Fourier Transform can be used. Although any orthogonal or any cyclic code can be used to practice the invention, cyclic codes are preferred because they are easier to generate. 
     The resulting real and imaginary component linear arrays of the results or chips out array are stored in a memory within the CDMA Mux  408  which is not separately shown. The components of these two arrays are then output on separate I and Q buses to a modulator  410  where they are used to amplitude modulate the amplitudes of two RF carriers that are 90 degrees out of phase using a Trellis modulation scheme in one embodiment or are used to control modulation in a carrierless modulation scheme described elsewhere herein. The resulting two AM carriers are summed and output on the transmission media  412 . This is done as illustrated in FIG. 23 in one embodiment. An up conversion or down conversion frequency translator (not shown in FIG. 23,  84  in FIG. 8) is used to move the resulting signal in frequency to the band designated for use. The frequency band designated for use depends upon whether the transmission media  12  is a cable TV system, satellite system etc. and further depends upon whether the signals are travelling in the upstream or downstream direction. 
     Referring to FIG. 23, more details of the coordination of the multiplexer  408  and the modulator  410  and the internal details of one embodiment of the modulator  410  in FIG. 8 are illustrated for the transmitter modulators in either the RU or CU. The result or chips out array is stored in memory  411  which is part of the CDMA MUX, and comprises the real or inphase array  409  and the imaginary or quadrature array  413  of the 144 result points or chips in the result space. On every chip clock, one result point or chip comprising a real component and an imaginary component is output on bus  451  to a bit parsing unit or bit splitter  453 . The bit parsing unit  453  splits off the real component and outputs those bits on bus  417 . The imaginary component will be parsed out, and those bits will be output on bus  419 . 
     Because the RF signals that carry the information from the 144 channels must share the transmission media with other RF signals having adjacent frequencies, two optional digital passband Nyquist shaping filters  421  and  423  are used to limit the bandwidth of the signals on buses  417  and  419  to 6 Mhz to avoid interference with signals on neighboring frequencies. The digital signals on buses  417  and  419 , when converted to their decimal equivalents usually have rapid transitions between levels in adjacent intervals. This is illustrated in FIG. 24 which is a plot of the changes in amplitude over time of the real components of the results vector for the array  409 . These filters  421  and  423  are Nyquist passband filters having center frequencies at the carrier frequency and having 6 dB bandwidth points which are each separated in frequency from the center frequency by a frequency gap 1/(2T c ) where T c  is the chip rate period, i.e., the time between transitions from one chip level to the other. The Nyquist filters  421  and  423  remove high frequency Fourier components caused by sharp edges in such signals. This filtering effectively rounds off corners of the waveform defined by the transitions between successive chip levels in the “chips out” array and limits most of the power density in the Fourier spectrum of such signals to a 6 Mhz band centered around the frequency of the RF carrier generated by local oscillator  425 . This local oscillator  425  generates a sine wave, RF carrier at a frequency selected to be compatible with the switching rate of CDMA multiplexer  408  and to not interfere with existing cable TV service signals on adjacent frequencies. Since, in one embodiment, the local oscillators in the RUs and CU that are used for the modulators and demodulators all run synchronously locked in phase to each other and to the phase of the master clock and master carrier signals used by the CU to transmit downstream data, and are kept in phase in the RUs by the carrier and clock recovery circuits described elsewhere herein, all the local oscillators that generate carriers will be designated  425  even though they are separate circuits one of which is in the CU and some of which are in the RUs. 
     The modulator uses a local oscillator COS wave carrier signal from the master carrier synthesizer in the case of the CU transmitter. In the case of the RU transmitter, the carrier signal comes from the tracking loop carrier recovery circuit ( 515  in FIG. 8) in the RU. In the preferred embodiment, the RU generates a local carrier signal which is phase coherent with the master carrier by using the frame detector and tracking loop circuitry previously discussed to synchronize the RU local clock with the master clock signal embedded in the CU Barker code. This RU local clock signal is then multiplied in a PLL to generate an RU local carrier reference signal which is phase coherent with the master carrier. The CU generates its master carrier in the same way using the master clock signal. However, the local carrier signal is generated, it is applied to the carrier input  427  of an amplitude modulator  429  which also receives the filtered real component of each chip on bus  431 . The modulator  429  modifies the amplitude of the carrier signal on line  427  in accordance with the amplitude of the decimal equivalent the real component on bus  431  and outputs the result on bus  443 . 
     The imaginary or quadrature component of each chip, after filtering, is input on bus  433  to another amplitude modulator  435 . This modulator receives at a carrier input  437  a sine wave of the same frequency as the cosine wave on line  427 , but shifted in phase by 90 degrees by phase shifter  439 . In an alternative embodiment, these local oscillator SIN and COS signals on lines  427  and  437  are actually generated in the carrier recovery circuit  515  in FIG.  8  and are locked in frequency and phase to the pilot channel tone sent downstream from the CU during timeslot  0 . Modulator  435  modifies the amplitude of the sine wave in accordance with the amplitude of the imaginary component on bus  433 , and outputs the result on line  441 . Lines  441  and  443  are coupled to a summer  445  which sums the two waveforms and outputs them on the shared transmission media via line  412 . 
     In some embodiments, the line  412  may be coupled to suitable interface circuitry to drive the signal on line  412  into a wireless or cellular system, a terrestrial microwave link, a coaxial cable of a cable TV, telephone or other system, a fiber optic link of a cable TV, telephone or other system, a local area or wide area network or any other media developed in the future for real time communication of data. Such interface circuitry is known and will not be described further herein. 
     As mentioned briefly above, in an alternative embodiment for purposes of carrier recovery by the RUs for downstream data, the master carrier signal is sent downstream as pilot channel data on a specific dedicated timeslot using a dedicated code. Referring to FIG. 28 there is shown a block diagram of the CU transceiver. The master carrier signal is applied to the CU transmitter modulator  410  via line  26 . The master carrier signal is generated by the master carrier synthesizer  28  from the master clock signal on line  22 . The pilot channel data in the CU transmitter is supplied to the forward error correction encoder on line  501  via a command and control buffer  503 . This pilot channel data is only transmitted downstream by the CU in the preferred embodiment, and the RUs do not transmit pilot channel data upstream since the CU knows the RU carrier is the same as the master carrier but somewhat offset in phase. In general, the command and control buffer stores data to be transmitted on the command and control channels for system management, contention resolution, ranging etc. by either the RU or CU transceiver. This other command and control data is received from the CPU  405  via bus  497 . Bus  505  couples this command and control data to an input of a switch  507  which has a second input coupled to receive the payload data on bus  360  from the framer. The switch selects one of these buses as the source of data which is output on bus  509  to the forward error correction encoder  402  for Trellis encoding. Switching of switch  507  is controlled by CPU  405  by a control signal on line  511  in some embodiments and is switched automatically by logic circuitry that knows when the command and control timeslots occur and when the payload timeslots occur. 
     In the transceiver of FIG. 8 for the RU, no pilot channel data is input to the command and control buffer. Instead the local carrier oscillator is either synchronized to the frequency and phase of the pilot channel or is synthesized from the recovered master clock. In embodiments where carrier recovery is performed, a local carrier oscillator inside carrier recovery circuit  515  is synchronized to the frequency and phase of the pilot channel signal broadcast in timeslot  0  from the CU. Such is the function of carrier recovery circuit  515  in FIG.  8 . The local reference carrier signal on line  427  is generated by using a tracking loop to lock the phase of a local carrier oscillator  425  in carrier recovery circuit  515  to the phase of the master carrier signal recovered from the pilot channel data (local carrier oscillator  425  can be located in the carrier recovery circuit or in the modulator  410 , but is preferably in the carrier recovery circuit). The local carrier reference is supplied to demodulator  460  in the RU receiver section as the COS signal on line  427  in embodiments having coherent detectors. In embodiments having rotational amplifiers, the demodulator/detector can be incoherent. In the CU receiver where a rotational amplifier is used, the demodulator  460  in FIG. 28 receives the synthesized master carrier signal on line  26  for incoherent demodulation. 
     Likewise, in the RU transmitter  401  of FIG. 8, the carrier recovery circuit  515  transmits to the modulator  410  a local oscillator signal on line  427  which is synchronized in frequency and phase to the pilot channel signal received by the RU receiver. This signal is input to the RU transmitter modulator so that its signals can be recovered by the CU receiver using the master carrier signal generated at the CU without the need for a carrier recovery tracking loop in the CU receiver to constantly track the RU transmitter&#39;s carrier. However, preamble data must be inserted into every RU timeslot&#39;s data for use by the CU receiver to periodically correct for the phase and amplitude errors of the carrier signals for that timeslot and that RU because every RU is at a different distance from the CU. Thus, even though the RU transmitter modulators uses a local carrier reference which is locked in frequency and phase to the master carrier at the CU, the differing propagation times and channel impairments cause phase and amplitude changes which are different for each RU. These phase and amplitude changes must be resolved by the CU separately for each RU so that the CU can adjust the phase of its master clock and master carrier signals for use in the CU receiver section when timeslot data from each particular RU is being received. This is the purpose of the preamble data which each RU transmits. The exact manner in which this is done will be described further below after completing the description of the carrier recovery circuit  515 . 
     The RU carrier recovery circuit  515  can be any conventional phase-locked loop clock recovery circuit, Mth power loop, Costas loop, suppressed carrier-tracking loop, etc . . . In the preferred embodiment, the carrier recovery circuit in the RU receivers takes the form shown in FIG.  25 . This circuit is basically a phase lock loop that compares a slicer error signal during timeslot  0  to the local oscillator frequency and phase generated by the voltage controlled oscillator  425 . The circuit then generates an error signal based upon the comparison to adjust the frequency and phase of a voltage controlled oscillator  425  to the frequency and phase of the pilot channel signal transmitted during timeslot  0 . The voltage controlled oscillator  425  serves as the local carrier reference signal for the demodulator  460  in the RU receiver section and the modulator  410  in the RU transmitter section. Specifically, the slicer detector  466  generates a slicer error signal on bus  519  which indicates at least the phase error between the received signal and a legitimate point in the constellation. 
     The error computing circuit  521  also receives a timeslot number enable signal on line  531  in FIGS. 25 and 8 from the CPU  405 . This signal indicates when timeslot  0  data is being received at slicer  466 , and causes the error computing circuit  521  to activate only when timeslot  0  pilot channel data is being received. Thus, carrier recovery and phase correction at the RU in this particular embodiment is an occasional rather than continuous function. During timeslot  0  the signal on  519  (which has been demodulated using the local oscillator signal on line  427 ) will indicate the phase error between the local oscillator signal on line  427  and the master carrier information in the pilot channel data. This phase error may be caused by a phase error between the local oscillator signal on line  427  and the master carrier pilot channel signal, or because of impairments on the channel such as noise, or it may be a combination of the two. Because the effect of noise is random but a phase error between the pilot channel and the local oscillator is constant until corrected, the phase error component caused by noise is removed by averaging in a low pass filter  523 . The slicer error signal on line  519  is coupled to an error computing circuit  521  which also receives the local oscillator signal on line  513 A which is coupled to line  427 . The phase error is calculated and output on line  525  to low pass filter  523  which averages the phase error over time thereby removing the noise component. The resulting average error signal is coupled on bus  527  to the error signal input of a voltage controlled oscillator  425  to generate the local carrier reference signal on line  427 . 
     An alternative carrier recovery arrangement is shown in FIG. 26 where elements that have like reference numbers to elements in FIG. 8 serve the same purpose in the combination and will not be discussed here. The embodiment of FIG. 26 uses an additional CDMA demultiplexer  461  which recovers only the pilot channel data on timeslot  0  by reversing the CDMA spreading process via a transpose matrix for the dedicated CDMA code used to spread timeslot  0 . The received timeslot  0  data is output on bus  465  to another slicer  463  in addition to the slicer  466  which compares the pilot channel data to a known point in the BPSK constellation used to transmit the pilot channel signal and develops a timeslot  0  slicer error signal which is output on line  519 . The slicer error signal is compared to the local oscillator signal on line  427  by an error compute circuit  521  and a phase error signal is output on line  531 . This phase error signal is averaged by low pass filter  523 , and the resulting error signal is coupled to the error signal input of the voltage controlled oscillator  425 . The output signal from the VCXO  425  is coupled via line  427  as the COS signal to the demodulator  460  and the modulator  410 . A 90 degree phase shift is applied to the COS signal in each one of these units to generate the SIN signal on line  437 . The SIN and COS signals can be in either digital or analog form in various species within the genus of the invention. 
     Referring to FIG. 28 showing the CU transceiver block diagram, the apparatus and method by which upstream carrier recovery, gain control and clock synchronization is achieved will be described. Even though all RU local oscillators are synchronized in frequency and phase with the master carrier information in the pilot channel data from the CU, the differing distances from each RU to the CU cause two different problems. The first is a different phase shift for the clock and carrier signals of each RU at the CU. The QAM signal demodulation used in the preferred embodiment depends for its accuracy on the ability to accurately distinguish between the amplitudes and phases of each received constellation point. The differing propagation times and differing channel impairments experienced by each RU&#39;s signal, cause both amplitude and phase errors in the received data that must be determined and corrected for to obtain accurate QAM demodulation at the CU receiver. 
     To correct for each RU&#39;s phase and amplitude errors, the CU must determine the phase error and amplitude error for each RU and correct for them individually as the timeslot data for each RU is being received. The way this is done is for each RU to send known preamble data to the CU in the timeslots currently assigned to that RU before the block of payload data is sent. This is done each time the RU transitions between an idle state when no data is being sent and an active state when upstream data is being sent. The CPU in the CU assigns the timeslots to the various RUs and so informs them in management and control messages on the management and control channels. This is done in response to bandwidth requests from the RU. The process by which the CU adjusts the phase of the master clock and master carrier signals for each RU is shown in FIG.  27 . 
     Upstream Carrier Recovery Error Correction Factor Per Timeslot 
     Referring to FIG. 27, there is shown a flow chart symbolizing the startup processing by the RU and the CU to determine the phase error in the clock and carrier signals for the RU each time the RU transitions from the idle to the active state. The first step in the sequence of an RU coming online is symbolized by block  1500  where the RU performs the ranging process described above to achieve frame synchronization. Next, the RU, in step  1502  performs the training process described later herein to set the coefficients of its filters to achieve proper equalization. Next, the RU determines whether it has any payload data to send in step  1504 , and, if not, the RU stays in the idle state by transitioning along path  1505 . If the RU has payload data to send, step  1506  is performed where the RU requests bandwidth from the CU in a management and control message modulated by amplitude shift keying. Because the CU does not yet know the phase error of the RU, upstream management and control messages from the RUs to the CUs are sent on the access channel portion of the management and control channels using a modulation scheme which does not require phase information such as amplitude shift keying or any other modulation scheme which does not require phase synchronization by the CU to the RU carrier and clock. Modulator  410  in FIG. 8 is used in the RU transmitter to do the ASK modulation as well as QAM modulation of the payload data simply by picking the two points of the 16 point QAM constellation that correspond to the two points of the ASK constellation during transmission of upstream management and control messages. Downstream management and control messages are sent using the same modulation scheme as is used for the downstream payload data. 
     In response to the bandwidth request, in step  1508 , the CU sends a downstream management and control message to the RU awarding bandwidth in the form of assignment of one or more timeslots. The RU responds to the bandwidth assignment by sending known preamble data to the CU during the assigned timeslots. FIG. 28 shows a block diagram of the CU modem. All the items in FIG. 28 that have the same reference numbers as items in FIG. 8 server the same purpose in the combination. The specific differences in functions of various blocks in the CU modem will be described briefly below. 
     The CU knows what the preamble data is supposed to be and knows when the preamble data is being received by virtue of knowing when the timeslots assigned to the RU sending the preamble data are being received. When preamble data is being received, the CPU  405  in FIG. 28 activates the CU PREAMBLE signal on line  1086 . This causes the slicer  467  to begin an iterative process to reduce the slicer error to as low a value as possible. The slicer  467  in the CU functions differently in some respects than the slicer  466  in the RU. The slicer  467  in the CU includes a rotational amplifier and a G 2  amplifier and a control circuit whereas the slicer/detector  466  in the RU uses a rotational amplifier and G 2  amplifier if it has incoherent demodulation. When preamble data is being received, the slicer circuit sets initial values for an amplitude error and a phase error for use in detecting the preamble data. The initial amplitude error signal is used by the G 2  amplifier in slicer  467  to correct for amplitude errors, and the initial phase error is used by the rotational amplifier to correct for phase error. The known preamble data point  3 - j  is then compared to the received data point, and the error is sent to the control circuit. The control circuit examines the error and readjusts the amplitude and phase error values used by the G 2  and rotational amplifiers inside slicer  467 . This process is continued until the error is zero. The final amplitude and phase error correction factors are then stored in memory  796  in a memory location devoted to the particular RU which sent the preamble data. The process described above is symbolized by steps  1510  and  1512  of FIG.  27 . 
     Step  1514  in FIG. 27 represents the process of determining the timeslot that is currently being received, looking up the modem ID currently assigned to that timeslot, retrieving the appropriate phase and amplitude error correction factor for that modem and applying those correction factors to a rotational amplifier and G 2  amplifier inside the slicer/detector circuit  467  in FIG.  28 . Step  1516  represents the process carried out by the G 2  and rotational amplifiers in the slicer/detector circuit  467  in correcting the phase and amplitude of the received payload data signals as they are received using the correction factors for the particular modem that sent the payload data. 
     The CU modem of FIG. 28 includes a master clock oscillator or input for a master clock signal  24  from which the master clock signal on line  24  is distributed to all circuits that need it. A master carrier synthesizer  28  receives the master clock signal and generates a master carrier signal  28  therefrom. The master carrier signal is distributed on line  26  to the modulator  410  of the CU transmitter, the demodulator  460 , and to any other circuit that needs it. The slicer/detector circuit  467  knows which RU&#39;s signals are currently being received from memory  464  by virtue of RU ID data received on bus  83  from the CPU  405 . This data is generated by the CPU from the timeslot allocation table. The CPU is informed which timeslots are currently being received by signals on bus  85  from the orthogonal demultiplexer or other circuitry not shown which functions to reassemble ATM packets using the 9th bit cell delimiter codes in the manner described in the TER-004.1P parent patent application which is incorporated by reference herein. The CPU informs the control circuitry within slicer/detector  467  which RU&#39;s phase and amplitude correction factors to use by sending RU ID data on bus  83  to the slicer. 
     In the embodiment shown in FIG. 8, the CPUs in the RUs keeps track of and help control the process of breaking the payload data from their peripherals/user devices into 8 bit bytes, adding a 9th bit to support the higher level protocol and sending the 9-bit bytes during the assigned timeslots. Before the payload data is sent however, the CPU or timing logic (not shown in FIG. 8) in the RU activates a Preamble signal on line  1094  which controls switching by a multiplexer  1076 . This multiplexer receives the encoded I and Q information vector payload data on buses  1068 A and  1070 A at one input and predetermined, fixed preamble data I and Q values on buses  1078  and  1080  at another input. When the switching control signal on line  1074  is activated, multiplexer selects the data on buses  1078  and  1080  for coupling to buses  1068 B and  1070 B for storage in memory  406 . The preamble data on buses  1078  and  1080  define a known point  3 - j  in the QAM constellation. 
     Line  1074 , buses  1078  and  1080  and multiplexer  1076  are only present in the RU transmitters since the technique described here is used only in the upstream data to achieve proper synchronization, so these circuits are absent from the CU transceiver block diagram of FIG.  28 . 
     In the CU receiver shown in FIG. 28, the slicer detector  467  is responsible for comparing the received data to the known preamble constellation point during preamble data reception to determine the amplitude and phase errors. The received signal takes the form: 
     
       
           a*e   jø   *s ( t ) 
       
     
     where 
     s(t) is the desired signal; 
     a=the amplitude error caused by channel impairments and the near-far problem; and 
     e jø =the phase error caused by channel impairments and the near-far problem. 
     The slicer detector  467  in FIG. 28 encompasses several circuits shown in the more detailed block diagram of the CU receiver discussed later herein. The slicer detector  467  operates to perform an iterative process to converge on a multiplication factor having amplitude and phase components to multiply times the received signal so as to cancel the amplitude and phase error such that s(t) is detected as the constellation point  3 - j  without any slicer error. The amplitude and phase error coefficients in the multiplication factor which reduce the slicer error to 0 are then stored in memory  796  for use by the slicer in receiving the payload data for the timeslot(s) assigned to the RU for which the multiplication factor was stored. 
     Specifically, the job of the CU receiver slicer detector  467  is to determine the correct 1/a and e −jø  coefficients in a multiplication factor of the form: 
     
       
         (1/a)*e −jø   (5) 
       
     
     where 
     1/a is the gain correction coefficient to solve the near-far problem and correct for channel impairments; and 
     e −jø  is the phase error correction coefficient to solve the near-far problem and correct for channel impairments and get the CU synchronized with each individual RU despite differing path lengths and differing channel impairments between the CU and each RU. 
     The Near-far Problem 
     The near-far problem involves interference with reception of weak signals transmitted from a remote RU by strong signals transmitted by a near RU. In the prior art, this is often solved by time division multiplexing so that the two transmitters are never transmitting at the same time. In the SCDMA environment, this solution will not work since all RUs have to be able to transmit whenever they need to transmit if bandwidth is available. Therefore, in the SCDMA example described here, the amplitude levels of the signals transmitted by the RUs are controlled so that all signals arriving from the RUs at the CU should arrive at approximately the same amplitudes, and channel impairment effects are corrected by gain level adjustments in the CU receiver at a point before the baseband signal enters the slicer so as to minimize interpretation errors caused by amplitude errors (the G 2  gain adjustment amplifier and rotational amplifier in slicer/detector  467  are located so as to receive the received signals before they get to the slicer). For a discussion of the iterative process carried out by this circuitry during the preamble for each timeslot to establish the values for the amplitude and phase error correction coefficients for use in receiving the payload data for that timeslot, see the discussion of the cooperation of G 2  amplifier  788 , rotational amplifier  765 , slicer  800 , control loop  781  and memory  796  in the detailed block diagram of the CU receiver discussed below. 
     Thus coherent modulation and detection is used for both upstream and downstream transmissions, but the coherent detection may be accomplished using rotational amplifiers. 
     The pilot channel data on timeslot  0  is spread with a dedicated CDMA code in CDMA multiplexer  408  for transmission on the timeslot  0  management and control channel as the pilot channel data which encodes the CU master carrier. Use of a pilot channel signal on one of the command and control channels is only one of the possibilities for distributing carrier frequency and phase information. Other possibilities are transmission of any modulated waveform which can be detected by the RU receivers in which the CU carrier frequency and phase information is encoded in the modulation. Note also that the RU can use another carrier frequency than the master carrier so long as phase coherence can be achieved in the CU receiver between the RU carrier and the master carrier. 
     The form of carrier recovery described above is only one way of achieving a coherent system with only one master clock and master carrier synthesizer. In FIG. 8 a coherent demodulator is shown having the structure of FIG.  29 . In the preferred embodiment shown in the block diagram of the RU and CU receivers of FIGS. 30 and 31, respectively, incoherent detection could also be used using any of the well known incoherent detection apparatus in conjunction with rotational amplifiers. In the CU receiver of FIG. 31, incoherent demodulation and a rotational amplifier is used. Alternative embodiments for incoherent receiver technology is described in Haykin,  Communication Systems , at page 503-505 and is hereby specifically incorporated by reference herein. 
     Another form of synchronization that is required is symbol or chip clock synchronization. The receiver must know the instants in time when the modulation can change its states from the amplitude and phase of one chip to that of the next. That is, the RU and CU receivers must know the start time and finish time of each chip in order to decipher what that chip was. This allows the receiver to determine when to sample and when to quench its product integrator or other chip state detection circuitry for purposes of starting the chip decoding process. 
     Chip Clock Synchronization 
     Symbol synchronization in the context of the SCDMA example herein is recovery of the CU chip clock in each RU. In the preferred embodiment, recovery of the CU master chip clock and master carrier is done by synchronization circuitry including frame detector  882 , control loop  781 , VCXO  784  and time base  886 , VCXO  808  and frequency synthesizer  760 . The master chip clock is recovered from the Barker code transmitted by the CU. The master carrier is synthesized by synthesizer  760  from the CU master chip clock in the preferred embodiment. There are at least two different ways of generating the master carrier in the RUs discussed herein and there are other ways known in the prior art. Any one of these ways will suffice to achiever carrier synchronization. One way to generate a synchronous local master carrier in the RUs is to control the phase of VCXO  808  in FIG. 30 using slicer error on bus  798  generated by pilot channel data. The preferred way is to synthesize the master carrier from the recovered master chip clock. Both embodiments outside the methods of recovering a master carrier known in the prior art are represented by FIG.  30 . 
     In one embodiment represented by FIG. 30, the CU master chip clock is recovered by correlating in the frame detector in each RU a known Barker code transmitted during every gap by the CU, with the Barker code encoding the chip clock therein. Each RU uses a correlator with an early-late gate to detect the Barker code and get the RU&#39;s chip clock synchronized with the CU chip clock encoded in the Barker code. This process of chip clock synchronization is carried out by the frame detector  513  in FIG.  8  and frame detector  882  FIG.  30 . The frame detector  513  and the frame detector  882  each includes both coarse and fine tuning circuitry. The coarse tuning circuitry performs downstream frame synchronization by locating the gap in each CU frame transmission by finding a known Barker code transmitted by the CU in the gap. Time base circuit  886  in FIG.  8  and FIG. 30 helps the frame detector find the Cu frame gap by generation of a window signal on line  1031  in FIG.  8  and FIG.  30 . This window signal activates the RU frame detector and tells it the limits within which the CU gap is expected. In the preferred embodiment, the CPU  405  generates the window signal as a GAP_a signal on line  902  (not shown in FIGS. 30 or  8 ) and sends it to the frame detector via bus  902  in FIGS. 8 and 30. In the preferred embodiment, the CPU is informed by the frame detector  513  as to when the CU gap and Barker code have been found by a signal on line  902  in FIG.  8  and by a connection not shown in FIG.  30 . That information is given by the CPU to the time base circuit  886  in FIGS. 8 and 30 by a signal on bus  1350 . 
     The frame detector  513  in the RU receivers is only active during this window signal, so clock recovery in the RUs is actually periodic and not continuous in the SCDMA example given by FIGS. 8 and 30 although a tracking loop is still used in the RU. The RU clock tracking loop forming part of the RU synchronization circuitry in FIG. 8 is frame detector  513 , VCO  1030 , time base  886  and line  1031 . A low pass filter is present to filter out noise, but is not shown in FIG.  8 . In FIG. 30, the clock tracking loop is frame detector  882 , low pass filter  115 , control loop  781 , VCXO  784 , time base  886  and window signal  1031  and clock line  117 . 
     The time base circuit  886  in FIG. 8 is comprised of a series of cascaded counter stages that receive a high speed input clock that is phased locked by the clock steering signal from the frame detector (line  900  in FIG.  8  and FIG.  30 ). The cascaded counters generate the chip clock, frame clock, superframe clock and kiloframe clock signals. In FIGS. 30 and 31, the time base circuit  886  includes the circuits  369 ,  359  and  371  in FIG. 11 as well as other circuits to generate the window signal as described further below and circuits to generate the other signals on FIG. 10 as well as the F 0  and F 4  frame and superframe signals. The time base  886  in both the receiver and transmitter of each modem also include a chip counter and a frame counter as well as sampling registers which are used to correctly align the timebase with external signals. Once the time base is aligned to these external signals, all internal timing needs of the modems are served by the time bases so that they do not depend upon external signals for operation, but the external signals are monitored for loss or shift. In the case of the CU, the external signals to which the time base is aligned are the frame timing, timeslot timing and bit timing signals from the TDMA input stream to the CU transmitter. These signals are monitored by connections not shown in FIGS. 31 and 28. In the case of the RU, the external signals include the recovered master clock and the Frame and Kiloframe signals derived from the downstream data. 
     The time base circuit  886  provides these signals which include receive frame timing reference information to any circuit in the receiver or transmitter that needs this information such as the receiver&#39;s orthogonal demultiplexer  462  in FIG.  8  and the orthogonal code demultiplexer  766  in the receiver of FIGS. 30 and 31. The time base circuit also continually checks the position of the gap by sampling a gap detect signal from the frame detector on line  1092  in FIG.  30  and FIG. 8 over multiple frames so as maintain frame synchronization and know when frame synchronization has been lost. When the gap position is lost, the modem immediately attempts to resynchronize to the gap. 
     The orthogonal code multiplexers in the RU and CU transmitters also get receive frame timing reference signals, but these frame timing reference signals establish the boundaries of the CU&#39;s frame timing reference since each RU transmitter times its transmissions and other processing so that frames transmitted therefrom arrive at the CU coincident with the CU frame boundaries. And of course the CU transmitter needs to transmit its frames in synchronism with the CU frame boundaries. To that end, the receive frame timing reference signal generated by the frame detector  882  in FIG. 30 and 513 in FIG. 8 is sent to the modem&#39;s local CPU or other control circuit  405  via bidirectional bus  902  in FIG.  8  and via bus  883  and DMA memory  763  in FIG.  30 . The CPU or other control circuit  405  then uses this frame timing reference to set the timing of the transmit frame timing delay T d  on line  499  to the transmitter frame circuits  400  in FIG.  8  and via lines  499  and  532  in FIG. 33 to the framer circuit  508  in FIG. 33 (FIG. 33 is a block diagram of the preferred embodiment of the RU transmitter). 
     The fine tuning circuitry in the frame detectors  513  and  882 , in FIGS. 8 and 30, respectively, performs clock recovery for chip clock synchronization by using early-late gating techniques in conjunction with correlation to generate a clock steering tracking error signal on bus  900 . This signal corrects the phase of the output clock signal from a voltage controlled oscillator  784  in FIG.  30  and VCO  1030  in FIG.  8 . This output clock signals is used by time base generator  886  in the RU to generate a local chip clock signal which is synchronous with the master chip clock in the CU. This recovered master chip clock signal and other timing signals generated from it are distributed to various circuits in the RU modem transmitter and receiver that need it to keep processing synchronized with processing in the CU. 
     The coarse tuning circuitry in the frame detectors  513  and  882  cooperates with a software process running in CPU  405  and the window signal generated by the time base under control of the CPU to help the frame detector locate the CU frame gaps. This is done using control and timing signals on bus  902  on the CPU and the real and imaginary data components on bus  904  output by the demodulator  460  in FIG.  8  and the matched filter  761  in FIG.  30 . This gap location process is accomplished by continually moving the boundary of a sliding correlation window established by the signals on bus  1031  in FIGS.  8  and  30  until a correlation peak appears at the same time at least twice consecutively. How this works will be explained in more detail next. 
     Frame Detector 
     Referring to FIG. 30, there is shown a block diagram of the preferred form of a ranging detector which forms the heart of frame detector in each RU and is used in the CU for ranging detection of Barker codes. The frame detector circuit of FIG. 30 may be hereafter referred to as the ranging detector even though it has frame detection and chip clock synchronization functions as well. 
     The ranging detector has an acquisition mode and a tracking mode. In acquisition mode, it is simply trying to rapidly find a known Barker code arriving in the collection of signals on bus  904  in FIGS. 8 and 30. In the preferred embodiment, where the transmit data is passed through a raised squared cosine filter, bus  904  is coupled to the output of a matched filter having a transfer function which is the inverse of a raised squared cosine function. But in other embodiments, these two filters may be eliminated. Bus  904  carries data defining the real part of the received signal on line  906  and the imaginary or quadrature part of the received signal on line  908 . 
     In acquisition mode, the interest is in quickly finding the gap by correlating the incoming signals with the known Barker code, but this can be done by simply looking at the sequence of signs of signals received since the Barker code is a known, unique sequence of chips of differing signs but constant amplitude. The Barker code can be located effectively in tracking mode by looking at only the sequence of differing signs in the received data. Therefore, in tracking mode, the CPU sends selection control signal ACO on bus  902  to control the state of switches  906  and  908  so as to select the signals on buses  910  and  912 . The signals on buses  910  and  912  are the outputs of circuits  914  and  916  which serve to compare the incoming signals on bus  904  to zero and output a first number if the sign of the incoming chip is +and output a second number if the sign of the incoming chip is −. When acq is not asserted, the raw data on buses  918  and  920  is selected for passing through switches  906  and  908 . The acq signal also passes through OR gate  922  to gate the output signals from switches  906  and  908  through to finite impulse response filters  924  and  926  in acquisition mode for correlation. The OR gate  922  also receives a GAP_a signal which is asserted by the CPU via bus  902  when the CPU thinks it is in the gap by virtue of signals from the frame detector. Therefore, the signals on buses  928  and  930  from switches  906  and  908  will be correlated by FIR filters all the time when the ranging detector is in acquisition mode and, while in tracking mode, only during the gap. 
     The FIR filters  924  and  926  have impulse response functions which are programmable and are set by the CPU  405  to match the Barker sequence which the receiver is looking for. The Barker sequence being sought is defined by data written by CPU  405  into register  932 . When this exact sequence of + and − chips resides in either one of the FIR filters, the filter output will peak. Absolute value circuits  934  and  936  are coupled to the outputs of the FIR filters, and output the absolute values of the FIR output signals on buses  938  and  940 . Circuit  942  has two different modes which are selected by the acq signal on line  943 . In acquisition mode when the receiver is trying to initially locate the gap, circuit  942  selects the greater of the signals on buses  946  or  948  for output on bus  944 . In tracking mode, the sum of the signals on buses  946  and  948  is output on bus  944 . 
     Comparator  950  acts to set a minimum threshold above which the FIR output peaks must rise before they are counted as possible reception of the CU Barker code. Comparator  950  compares the signals on bus  944  to a threshold level on bus  945 , and, if the threshold is exceeded, outputs a logic 1 on bus  951  during the interval when the threshold is exceeded. The threshold level is set by data written into register  952  by CPU  405  via bus  902  (bus  902  contains more signals lines than just the two lines shown in FIG.  30 ). The number of peaks is counted by a false alarm counter  952  the output of which is stored in register  960  which is periodically read by the CPU in a process of monitoring and controlling the ranging detector. A process in CPU  405  which monitors the number of false alarms, sets the number of frames over which false alarms will be counted by writing a number of frames into register  956 . This number is loaded into interval counter  954  which counts down from that number by counting the GAP_b signals on line  957  which occur one per frame. When the count reaches zero, line  958  is activated which clears the false alarm counter  952 , strobes the count before clearing into register  960  and reloads counter  954  from register  956 . When the CPU determines that the number of false peaks is too large according to the number in register  960 , it raises the threshold by writing new data to register  952  to raise the threshold. 
     Course tuning to find the gap is accomplished by the ranging detector as follows. The CPU starts with an estimate of when it thinks the gap will start. At that time, signal GAP_a on bus  902  is asserted during each frame interval. The CPU only wants to look at peaks during the gap in each frame interval, so it uses a sliding window to restrict the time during which it is looking for peaks. The sliding window is symbolized by bracket  962  in FIG.  35 . The boundaries of this window are established by data written by CPU  405  to register  964  in a manner to be described below. 
     Circuit  970  passes only the first peak on the output of the AND gate  968  which occurs after the GAP_a signal indicates the gap is thought to have started. A time base counter  972  counts chip clock signals on line  974  and is cleared by the GAP_a signal every frame. When circuit  970  passes a peak (actually a logic 1 level) through on bus  976 , the current count of the time base counter  972  output on bus  980  is sampled and stored in register  978 . The count value on bus  980  is also coupled to a comparison input of a greater than or equal to comparator  965 , the other input of which is coupled to receive the output of the register  964 . The output of the comparator  965  is the gating signal on line  966 . Since the count of time base counter  972  will be reset to 0 at the moment the CPU thinks the gap is starting, the count stored in register  978  represents an offset error indicating how much later the gap may have actually started compared to the time the CPU thought the gap was starting. 
     FIG. 35 is a timing diagram that helps explain the course tuning process to find the time the CU frame gap occurs which is carried out by the RU receivers. Timeline A of FIG. 35 represents the initial sliding window position  962  set by the CPU during a first frame before it is sure where the gap is and shows the times of two peaks observed during frame  1 . Timeline B represents the position of the sliding window and the peaks observed during frame  2 . Initially, the CPU does not know where the gap is, so the software process decides to watch for peaks on line  976  for the whole frame. Accordingly, the CPU writes a 0 into register  964  at time T 0  and simultaneously activates the GAP_a signal. Activation of the GAP_a signal resets the timebase counter  972  and drives a logic 0 onto bus  980 . The 0 in register  964  is compared to the 0 on bus  980  by greater than or equal to comparator  965  which finds an equality and sets line  966  to logic 1 thereby gating pulses on bus  951  from the threshold comparator through to the first pulse selection circuit  970 . Comparator  965  drives line  966  to logic 1 anytime the number on bus  980  is greater than or equal to the output of register  964 . This action opens sliding pulse observation window  962  in FIG. 35 at time T 0 . The window will remain open until the end of the frame. 
     During frame  1 , shown on timeline A of FIG. 35, a noise pulse  990  is gated through circuit  970  at time T 1 , and the actual Barker code pulse  992 A which occurs at time T 7  is blocked by circuit  970 . The occurrence of noise pulse  990  causes sampling of the count on bus  980  by the register  978 , which is indicated in FIG. 35 as sample  1  at time T 1 . This value is read by the gap acquisition process executing on CPU  405  and stored for later comparison. 
     Because the noise pulse  990  was random, it does not occur again at time T 1  in the second frame shown on timeline B of FIG.  35 . Instead, another noise pulse  994  occurs at time T 3 , later than T 1 , and another Barker code pulse  992 B occurs at time T 7 . First pulse selection circuit again gates pulse  994  through and blocks pulse  992 B. This causes the taking of sample  2  of the count on bus  980  during frame  2 . The coarse tuning gap acquisition process reads the value stored in register  978  and compares this value to the value previously read from this register during frame  1 . The CPU concludes pulse  990  occurred at a different time than pulse  994 , and, therefore, pulse  990  was noise and cannot be attributed to the Barker code because if it were the Barker code, it would not be random and would have occurred at the same time. Accordingly, the gap acquisition process moves the position of the window  962  for frame  3  to open at a time just before the occurrence of pulse  994  so as to eliminate any pulses before that time from consideration but so as to analyze pulse  994  to see if it is attributable to the Barker code. The CPU gap acquisition process moves the position of window  962  by taking the sample  2  number from register  978 , subtracting a fixed amount from it, and writing the result to register  964 . 
     The situation for frame  3  is shown on timeline C of FIG.  35 . The window  962  opens at time T 2 , but because pulse  994  in frame  2  was noise, it does not occur again in frame  3  at time T 3 . Instead, noise pulse  996  occurs at time T 5 , and is gated through by circuit  970  while the actual Barker code pulse  992 C is blocked. Pulse  996  causes sample  3  to be taken. The gap acquisition process compares sample  3  to sample  2  and concludes that pulse  994  was noise because pulse  996  did not occur at the same relative time (relative to the occurrence of GAP_a). Accordingly, the gap acquisition process concludes that the window  962  can be moved again. This time, the window is moved to open at a time T 4  just before the time of occurrence of pulse  996  at time T 5 . 
     During frame  4 , window  962  opens at time T 4 , but no pulse occurs again at relative time T 5 , but the Barker code pulse  992 D occurs again at time T 7 . This Barker code pulse is gated through by circuit  970  and causes sample  4  to be taken. The gap acquisition process reads sample  4  and compares it to sample  3 , and decides that pulse  996  was noise because pulse  992 D did not occur at the same relative time. Accordingly, the gap acquisition process moves the position of window  962  again so as to open at a time T 6  just before the occurrence of pulse  992 D. 
     The situation during frame  5  is shown on timeline E of FIG.  35 . The window opens at time T 6  thereby precluding consideration of any pulses occurring before T 6 . Another Barker code pulse  992 E occurs again at relative time T 7  which is gated through as the first pulse in this frame after the window opened by circuit  970 . This causes the taking of sample  5  which the gap acquisition process compares to sample  4  and concludes that the relative times of occurrence of pulses  992 D and  992 E were the same. The gap acquisition process then concludes that pulses  992 D and  992 E were Barker code pulses and that it has found the gap. Accordingly, the gap acquisition process leaves the window  962  set to open at time T 6  in frame  6  shown on timeline F of FIG. 35 thereby ignoring noise pulses  998  and  1000  which occur before T 6 . The gap acquisition process then moves the time of activation of GAP_a to time T 7 , as shown on timeline G in FIG. 35, and switches the ranging detector to go into tracking mode for the chip clock recovery process by de-asserting the acq signal on bus  902 . 
     The chip clock recovery process is carried out by early-late gate sampling circuitry in FIG. 30 and, in the preferred embodiment, begins after the gap acquisition process. The basic concept is illustrated in FIG. 36 which is a diagram of the sampling by the early-late gating circuitry of the output of the FIR filters (correlator output) when phase lock with the chip clock has been achieved. Curve  1002  represents the output signal on bus  944  from the correlation process that occurs in the FIR filters  924  and  926  between the known Barker code (defined by coefficients in register  932 ) and the incoming signal. The major peak  1004  centered on time T 0  (a different T 0  than in FIG. 35) represents the correlator output when the Barker code sent in the gap by the CU arrives and is perfectly aligned in the FIR filters  924  and  926  with the data in the register  932 . This register contains data defining the + and − polarity sequence of the individual elements of the Barker code sent by the CU. Every CT-2 chip clock (8 chip clocks), a new digital sample of the received signal enters the FIR filters. The FIR filters do a summation of the results of each stage every CT-2 chip clock. When all the samples of the Barker code have entered the FIR and are aligned with the + and − polarity sequence that defines the Barker code the receiver is looking for, the summation on the CT-2 chip clock that results in the alignment causes the peak  1004  at the output on line  944 . Peaks  1006  and  1008  are examples of the summation results in the FIR filter before and after perfect alignment occurs. Points  1010  and  1012  represent sample points each of which is spaced apart from time T 0  by one CT-2 chip clock. When the local clock oscillator  784  in the embodiment of FIG. 30 or VCO  1030  in FIG. 8 is exactly aligned in phase with the phase of the master clock signal generated by the CU, the amplitudes of sample points at  1010  and  1012  will be the same. When there is some phase error, the two sample point  1010  and  1012  will have unequal amplitudes because pulse  1004  will not be symmetrically centered on T 0 . This generates the error signal CLOCK STEERING on line  900  in FIGS. 8 and 30 which causes the phase of a chip clock voltage controlled oscillator in the phase locked loop to shift in such a manner as to alter the timing in which the data samples are fed into the FIR filters  924  and  926  so as to get the correlator main pulse  1004  to center on time T 0 . 
     The manner in which this clock recovery process is carried out by the circuitry of FIG. 30 is as follows. Circuits  1014  and  1016  are the digital equivalents of sample and hold circuits. Circuits  1018  and  1020  are each delay circuits that each impose a CT-2 chip clock delay on a sample signal on line  1022 . This sample signal is generated by the CPU  405  once per frame at a predetermined time in the gap after the GAP_a signal is activated. The sample signal cause circuit  1014  to sample the magnitude of the pulse  1004  on line  944  so as to take sample  1010  in FIG.  36 . This sample value is coupled to one input of a subtractor  1024 , the other input of which is the magnitude of the signal on bus  944  (all processing is digital in the preferred embodiment). The subtractor  1024  constantly subtracts the first sample value  1010  stored in register  1014  from the changing values on bus  944  and presents the difference on bus  1026 . Two CT-2 chip clocks later, the sample signal on line  1022  reaches register  1016  and causes it to store the difference value at that time on bus  1026 . The value stored in register  1016  is the difference in amplitude between samples  1010  and  1012  in FIG.  36 . This value is the track error signal on bus  900 . The CLOCK STEERING signal on line  900  is digitally integrated in a low pass filter (not shown in FIG.  8 —block  115  in FIG. 30) to eliminate the effect of random noise, and the result is used as an error signal to correct the phase of a voltage controlled oscillator  784  in FIG.  30  and VCO  1030  in FIG.  8 . These voltage controlled oscillators serve to generate the local chip clock reference signals in the embodiments of FIGS. 30 and 8. This chip clock reference signal is coupled on bus  1032  in FIG.  8  and buses  786 ,  793  and  888  in FIG. 30 to time base  886  which generates the other timing signals needed to synchronize operations of the receiver and transmitter in FIG.  8 . 
     In alternative embodiments, the chip clock could be recovered by transmitting the chip clock with the data bearing signal in multiplexed form and then using appropriate filtering or demultiplexing at the RU to extract the chip clock. Another possibility is to use a noncoherent detector to extract the chip clock taking advantage of the fact that the chip clock timing is more stable than the carrier phase. The carrier is then recovered by processing the detector output during every clocked interval. Another possibility where clock recovery follows carrier recovery, as is done in the preferred embodiment, is to extract the chip clock from demodulated baseband output from the CDMA demultiplexer. 
     In addition, all the RUs may possibly synchronize to a single common external time source such as GPS satellite time information although synchronization to within 1 microsecond may not be adequate accuracy in all applications. Any conventional methodology for achieving synchronization of the RU chip clocks and local oscillator signals to the corresponding signals in the CU will suffice for purposes of practicing the invention. 
     The ranging detector of FIG. 30 also includes circuitry to determine when a Barker code is exactly centered in the gap. This capability is used in the CU version of the ranging detector during the fine tuning process at the end of the ranging process where the CU sends instructions to the RU on how to adjust its transmit frame timing delay to exactly center its Barker code in the gap. How this is done will be explained with reference to FIG. 37 which illustrates the 3 permissible patterns of data at the output of comparator  950  for a centered Barker code condition to be declared. Basically, the gap is 32 chip clocks wide, and is represented by window  1034 . Comparator  950  will output 32 logic 0s or 1s during the gap interval, and these are shifted into shift register  1036 . Two latches  1038  and  1040 , each 16 bits wide, have their inputs coupled to the 32 bit parallel output bus  1042  of the shift register. These two registers  1038  and  1040  are constantly enabled, and are loaded with the contents on bus  1042  at the end of the gap with one taking the lower 16 bits and the other taking the upper 16 bits. For the Barker code to be centered only the three bit patterns shown in FIG. 37 are permissible. The first bit pattern on line A indicates two logic 1 s on either side of the gap centerline  1044  and represents the data pattern that will be present in latches  1038  and  1040  when the RU&#39;s transmitted Barker code has been exactly centered. The bit patterns on lines B and C represent acceptable conditions where the Barker code is not exactly centered. The data patterns in registers  1038  and  1040  are read by the ranging process in execution on CPU  405  during the fine tuning process to deduce what instructions to give the RU to change its transmit frame timing delay T d  so as to move its Barker code toward the center of the gap. 
     Returning to the consideration of FIG. 8, the remaining receiver side circuitry of the transceiver will be described in more detail. As is the case with the transmit channel, the processing performed in the receiver may be performed using analog or digital or some combination of analog and digital circuitry. The receiver will be described as if all processing was digital as it is in the preferred embodiment. The signal received from the shared transmission media  412  is passed through an analog-to-digital converter (not shown) and the resulting digital data stream is passed to a demodulator  460 . 
     FIG. 29 is a more detailed diagram of the structure of the demodulator  460  in the receiver. The received analog signal from the shared transmission media is coupled on line  461  to the analog input of an A/D converter  463 . The stream of digital data resulting from the analog-to-digital conversion is simultaneously fed to two multipliers  465  and  467 . Multiplier  465  receives as its other input on line  481 , a stream of digital values that define the master carrier in the CU or the local carrier reference in the RU having the same frequency and synchronous in phase with the RF carrier sine wave on line  427  in FIG.  8 . Multiplier  467  receives as its other input on line  427 , a cosine signal which is synchronous with the CU&#39;s master carrier pilot channel broadcast in timeslot  0  but 90 degrees out of phase therewith. The inputs labelled SIN and COS in FIG. 26 are generated by the carrier recovery circuit  515  in the embodiment of FIG. 8 where carrier recovery is performed. The code dedicated to the pilot channel is used to spread the pilot channel signal using conventional spread spectrum techniques. Each receiver decodes the pilot channel using this same code to recover the pilot channel carrier signal and applies the recovered signal to a phase detector in a phase lock loop which is used as a local oscillator source for the demodulator in each RU receiver section and the modulator in the RU transmitter section. 
     The results output from the demodulator on lines  469  and  471  are digital baseband data streams which basically defines the mix products comprised of a fundamental carrier frequency and upper and lower sidebands. Digital filters  473  and  475  filter out the desired sidebands that contain the real and imaginary parts of each chip or result point that was transmitted. The stream of quadrature or imaginary components of the received chips are output on bus  477 . The stream of inphase or real components of the received chips are output on bus  479 . Per the teachings of the invention, the recovered clock and carrier signals in the RU are then used for transmissions by the RU to the CU so that the CU can coherently communicate with the RU&#39;s without having to synchronize to different clock and carrier signals used by the RU&#39;s. 
     In alternative embodiments, the RUs can use their own clock and carrier signals which are unrelated to the CU&#39;s versions and the CU can contain its own phase lock loop circuitry to recover these signals and synchronize to them in order to demodulate and interpret the data transmitted by the RUs. 
     In some embodiments, the streams of real and imaginary components of the  144  chips of each symbol on buses  477  and  479  are stored in two linear arrays in CDMA Demultiplexer  462  in FIG.  8 . The CDMA Demultiplexer  462  multiplies each of the real and imaginary component arrays times the transpose of the code matrix used by the CDMA MUX  408  of whatever RU or CU that transmitted the data to reverse the orthogonal code encoding process. This matrix multiplication process results in two linear arrays of decoded chip real and imaginary parts for each symbol. These arrays are stored by the CDMA Demultiplexer  462  in memory  464 . In alternative embodiments, the CDMA Demultiplexer processes the two streams of real and imaginary components “on the fly” such that they do not have to be first stored as input arrays in a memory in the CDMA Demultiplexer  462 . 
     The mapping by orthogonal code transformation from the constellation of possible input points shown in FIG. 21 leads to a constellation of possible points in a received chip space. A detector  466  of the RU shown in FIG. 8 or detector  467  of the CU shown in FIG. 28 examines the points in each of the arrays and compares the received chip points they define against the legitimate possible points in the received chip space. The detector, otherwise known as a slicer, is a known type of circuit and no further details are necessary herein. The function of the detector is to restore the gain and phase of the received signal using G 2  and rotational amplifiers, recover the pilot channel data therefrom and generate slicer error signals on bus  517  for the pilot channel data for use by carrier recovery circuit  515  in FIG. 8 so as to allow generation of a local carrier reference which is in synchronization with the master carrier, determine the boundaries of each payload data chip and determine the values for the I and Q coordinates of each received chip and compare the I and Q coordinates of each received chip point against the closest points in the constellation of legitimate possible points in the received chip space that could have been transmitted. The detector then makes a preliminary decision as to which of the possible legitimate points in the received chip constellation each received chip is likely to be. 
     The detector  466  outputs its preliminary determinations to a Viterbi Decoder  468  which performs the prior art Viterbi algorithm to determine the actual constellation point sent with each chip. The Viterbi Decoder uses the 4th bit in each chip of each symbol to detect and correct errors using the Viterbi algorithm to derive the most probable tribit path defined by the points actually sent from the path in the received chip space defined by the 4-bit components of the symbols actually received. The addition of the 4th bit to each tribit converts the input constellation from an 8 point to a 16 point constellation by addition of redundancy. The addition of this redundant 4th bit increases the distance between the path through a space defined by successive input constellations, one for each symbol time. The fact that the chip path is farther from the 3 bit path makes it easier for the receiver to divine from the noise corrupted received data what the actual tribits transmitted were. Viterbi Decoders are well known in the art of digital communications, and no further details will be given here. This Viterbi algorithm could be carried out by a programmed digital computer if slow speed is enough or by a dedicated hardware circuit if speed is important. Viterbi Decoder based systems are used by Qualcomm, Inc. in San Diego in cellular phone systems to combat noise in digital cellular phone transmissions, and the details of their patents and products are hereby incorporated by reference. 
     The output data points from the Viterbi Decoder are a stream of tribits. These tribits are stored in a memory in a deframer circuit  470  which functions to reassemble a replica of the TDMA data stream in the time domain from the incoming stream of chips or tribits comprising each symbol. This process is done by reversing the reading and writing processes described above in filling and emptying the framer memory  300  of FIG.  14 . 
     Fallback Mode 
     Fallback mode is entered when noise power gets too high. The noise power is detected by the CU, and when it reaches a predetermined threshold, the CU commands all RU modems to reduce the amount of payload in each symbol and add more redundancy. Fallback mode is implemented by a mode control signal on line  530  in FIG. 32 to the encoder circuit  526  (the connection to the CPU  405  is not shown). This mode control signal can command three modes: idle mode where the encoder pass the tribits adding only zeroes as the 4th bit; normal mode where 4th bits are added based upon the previous state for that timeslot during the last symbol time; and fallback mode where more redundant bits are added to each 4-bit group and correspondingly less payload data in included in each 4 bit group. 
     Code Diversity in CDMA to Improve Performance 
     Referring to FIG. 38, there is shown a diagram of a machine to achieve code diversity in CDMA systems so as to improve the performance thereof. The code diversity apparatus and processing described herein is useful in any digital data communication system wherein code diversity is used to keep separate conversations separate. It has been found by the applicants that in CDMA systems, some codes are more sensitive than others to misalignment and narrow band interference and will have higher bit error rates. In most systems, the higher bit error rate caused by one code would be unacceptable and the codes which are more sensitive to noise could not be used. In some systems with large numbers of channels of digital data to send, there are only one or a few code sets which have enough codes which are orthogonal to accommodate all the channels. For example, with 144 different timeslots/channels, there is only one code set with 144 orthogonal codes. Rather than omit the codes which are too sensitive and possibly not have enough codes to accommodate all channels, the codes are shuffled between channels randomly thereby spreading usage of the weaker codes around among the different channels. Code diversity requires coordination between code diversity tables in the RU transmitter and the CU receiver so that both are using the same codes during the same frames to encode and decode specific timeslot data. There are also restricted code lists that list codes that are not to be used. RUs that implement code diversity must maintain their code diversity and restricted code tables up to date with CU downstream messages to remain operational. The RU computer must download a checksum verified copy of the current tables and activate the tables before the modem can initiate or receive connection requests. Downstream messages updating the code diversity and restricted code list tables are sent from the CU with a superframe tag number which defines when the update is effective. Every downstream message includes a table checksum against which the RU modem can check its own checksum to insure validity of its tables. The CU broadcasts its checksum each superframe, and each RU maintains an independent checksum. 
     This code diversity concept can be used in any CDMA system. In CDMA systems where all the timeslot data is collected in one physical location, code diversity can be implemented using a shuffler  500  shown in FIG.  38 . In this application, the shuffler is a crossbar switch which receives a plurality of inputs  502  and has a plurality of outputs  504 . The inputs  502  each carry the digital data from one timeslot. The outputs  504  each carry the digital data from a randomly assigned one of the inputs, which changes periodically, and are coupled to matrix multiplication circuitry such that each timeslot&#39;s data gets multiplied by a different code during different periods. The inputs  502  are coupled to the inputs of a crossbar switch within shuffler  502  which periodically or randomly shuffles each of the inputs to a different output line for coupling to a multiplier for multiplication by a CDMA spreading code assigned to that output line. The crossbar switch can take the form of the high speed crossbar switch disclosed in U.S. Pat. No. 5,355,035 which is hereby incorporated by reference. 
     In systems like the CDMA CATV system disclosed herein where at each RU not all the timeslot data for all 144 timeslots is present at each location, the shuffler takes a different form and is located in the CU. In this embodiment, the inputs  502  represent requests for bandwidth relayed to the CU by all the RU&#39;s, and the outputs  504  represent code assignment transmissions to the RU&#39;s over the command and control channels where the code assignments could change every frame or even after transmission of each symbol. At the CU however, all the timeslot data of channels to be transmitted to the RU&#39;s is located in one place, so the shuffler can take the physical crossbar switch form previously discussed in the paragraph next above. The shuffler  500  can also take the form of a suitably programmed computer to shuffle the timeslots to different codes as well as perform the matrix multiplication. 
     The use of this shuffling technique spreads the weak codes around but the weak codes still cause errors. If the level of errors generated by this technique cannot be tolerated, forward error correction is used in conjunction with the code diversity to eliminate the errors. Forward error correction means sufficient redundant bits are inserted into the data stream by the encoder  526  in the CU and RU transmitters to allow any errors to be corrected without the need for retransmission of frames with errors. In the specific embodiments disclosed herein, Trellis modulation is used with a convolutional encoder in each RU and CU transmitter to calculate and add to each tribit a redundant 4th bit. These 4th bits are used by the receivers and Viterbi Decoders therein to correct errors by making judgments from the received data which points from the constellation of possible points were actually sent. 
     In the preferred embodiment for a transmitter described below with reference to FIG. 32, a diversity shuffler  506  implements code diversity by coordinating the shuffling of timeslot data to different, randomly selected CDMA spreading codes by the signals on buses  532  to the framer  508  and the signals on bus  533  to the buffer  533 . This will be described in more detail below. 
     Preferred RU Transmitter Block Diagram 
     Referring to FIG. 32, there is shown a block diagram of the preferred species of transmitter circuitry for an SCDMA species of CU transmitter within the genus of the invention. FIG. 33 is the preferred RU transmitter embodiment for a species within the genus of the invention. The transmitters of FIGS. 32 and 33 will be discussed jointly and only differences between them will be separately discussed. References to the transmitter should be understood as referring to either the RU or CU type. The transmitter is used in the transceivers of the RU and CU modems to transmit via synchronous CDMA data in both the upstream and downstream direction although transmission in the downstream direction can be by TDMA or any other scheme without adversely affecting performance. SCDMA is preferred for the upstream direction because of its increased throughput capacity. In the CU access control circuitry  540 , the data points for management and control information is chosen to be ASK or DQPSK points from the QAM constellation so that ranging communications and other communications that need to occur before the RU receiver achieves phase synchronization can still occur. 
     In FIG. 32, block  506  is the diversity code shuffler that implements the time to code transformation. The code shuffler receives a pseudorandom seed number on bus  499  which controls the pseudorandom order of shuffling of codes such that the various timeslots or channels are not always encoded with the same CDMA codes. Bus  499  also carries Tss data which defines which timeslots are assigned to this RU transmitter and an RU/CU signal which tells the code shuffler whether it is operating in an RU or CU. The R 1  data on bus  499  defines reserved codes which cannot be used, and the T d  data is received from the CPU and receiver frame detector circuitry to set the transmit frame timing delay value for this RU so as to hit the gap with its Barker code thereby achieving frame synchronization. 
     Block  508  is the framer circuitry that implements the variable transmit frame timing delays needed to implement the ranging process to achieve the necessary frame synchronization and time alignment of the CDMA spread channel data for synchronous CDMA. The framer circuitry  508  is described in more detail in FIG.  12 . Block  548  is a buffer that stores the shuffled 4 bit groups of symbol elements which serve as the information vector [b] for the matrix multiplication performed by the CDMA Multiplexer  527 . Code diversity can be implemented by block  506  by controlling the order of tribits read for each symbol from framer memory  508  via read pointers sent to the framer on bus  532 , and the framer structure must be such that read pointers can be externally supplied. The time delay value Td is supplied to the framer via bus  599 ′. The tribits exit the framer on bus  518  in the order dictated by the read pointers supplied either externally via bus  532  or internally generated. They are pseudorandomly scrambled by scrambler  524  in the manner described below (in the preferred embodiment) and redundant bits are added by encoder  526  if operating in normal or fallback mode. The randomizer machine scrambles the incoming data for privacy and to make the data look more like white noise. This reduces the dynamic range at the output of the transmitter. The randomizer receives its scrambling instructions from a scramble register  525  which receives and stores a seed code on bus  529 . In some embodiments, the randomizer  524  can be omitted. 
     Encoder  526  adds at least one bit to every tribit in the preferred embodiment to implement Trellis modulation. Some embodiments have no encoder, and some embodiments have an encoder which has no idle and/or no fallback mode. 
     The encoded bits are divided into real (or inphase) and imaginary groups by dividing each encoded tribit in half and outputting the first 2 bits as the real bits on bus  517   r  and the last two bits on bus  517   i . Buses  517   r  and  517   i  are coupled to a switching circuit  544  which also receives as inputs real and imaginary components of access channel information on buses  542   r  and  542   i . During normal payload transmission operations, switching circuit  544  selects the data on buses  517   r  and  517   i  for coupling on buses  546   r  and  546   i  to buffer memory  548 . During access channel operations, switching circuit  544 , under control of microprocessor  405  or other timing logic, selects the data on buses  542   r  and  542   i  for coupling on buses  546   r  and  546   i , respectively. The real and imaginary components in each tribit on buses  546   r  and  546   i  are written into buffer  548  in the order dictated by write addresses on bus  533 . Elsewhere herein, the manner in which the multiplexer  544  is operated to overlay media access control data on buses  542   r  and  542   i  with payload data on buses  517   r  and  517   i  in buffer  548  is described. Buffer  548 , when fully written, during each symbol time has 144 4-bit elements comprising an information vector the order of which is randomly scrambled anew each symbol time in the preferred embodiment. In other embodiments, the codes may be assigned sequentially during each symbol for all active timeslots, or a rolling sequential assignment of codes to all active timeslots may be used. 
     Referring to FIG. 39, there is shown a block diagram of a simple embodiment for the code diversity shuffler  506 . This embodiment does not do random shuffling but does a rolling shuffle in the following manner. Each RU and the CU has a code diversity shuffler of the same type and all shufflers operate synchronously to shuffle the same timeslots to the same codes simultaneously. A timeslot scanning counter  601  increments from 0 to 143 in synchronism with a system clock on line  603 . This count is output on bus  532  as an address to a random access memory  605  which stores a copy of the channel activity table. The channel activity table is a table which stores data indicating which of the  144  timeslots are currently being used. The CU broadcasts data to all RUs indicating which channels are currently assigned, and each RU updates its activity table using circuitry not shown in FIG.  39 . Bus  532  carrying the timeslot scanning counter output is also coupled to the framer  508 , and the count on bus  532  acts as a read pointer controlling which tribit from the current symbol being read is output from the framer on bus  518 . The count on bus  532  is also coupled to an address input of RAM  605  and causes data to be output on bus  607  indicating whether the channel corresponding to the current count is currently assigned. This data is, for example, a logic 1 if the timeslot is assigned and logic 0 if not. The bus  607  is coupled to the increment input of a timeslot activity counter  609  which has its clock input coupled to the system clock on line  603 . When a logic 1 is output on bus  607 , the timeslot activity counter  609  increments on the next upward clock transition. Counter  609  counts sequentially from 0 to 143 and then rolls back over to zero. The output of the counter  609  on bus  533  is coupled as a write pointer to the address input of buffer memory  548  in FIG.  32  and controls where the tribit output by the framer  508  is written, after encoding by encoder  526 , in the information vector [b] stored in buffer memory  548 . The read pointer on bus  532  is also coupled to a symbol count decoder  611  which generates an incrementation signal on line  613  each time the count on bus  532  reaches  143  thereby indicating the first tribit of a new symbol will be read on the next upward system clock transition. A symbol counter  615  then increments on the next upward clock transition to generate a new symbol count on bus  617 . This symbol count is coupled to a preset input of the timeslot activity counter  609  and causes the timeslot activity counter to be preset to whatever symbol count exists on bus  617  and to continue to increment from there as. active timeslots are found. When symbol counter reaches  143 , it rolls over to 0. Thus, for each new symbol, the timeslot activity counter starts incrementing from a new number. This causes a rolling shuffle of the positions in which the 4-bit groups are placed in buffer memory  548  thereby causing each active timeslot to be spread using a different code during each new symbol to achieve code diversity. 
     FIG. 40 is a block diagram of another embodiment for a code diversity shuffler that can be substituted for diversity shuffler  506  in FIG.  32 . This embodiment does a pseudorandom shuffle of codes using a shuffling table filled with pseudorandomly distributed write pointers. In FIG. 40, all elements are the same as in FIG. 39, except that the output on bus  533  from the timeslot activity counter  609  is coupled as an address input to a memory  619  which can be either a RAM, ROM, PROM, EEPROM or EPROM. Memory  619  stores a collection of 144 write pointers which are pseudorandomly distributed relative to the sequential address inputs. Each count on bus  607  from the timeslot activity counter causes whatever pseudorandom write pointer is stored in that address in memory  619  to be output as the write pointer on bus  533  to buffer memory  548  in FIG.  32 . All RUs and CUs have an identical copy of the pseudorandom shuffle table stored in memory  619 , and all RU&#39;s and the CU synchronously scan the activity table and synchronously, pseudorandomly assign the same CDMA spreading codes to the active timeslots. 
     FIG. 41 shows a block diagram of a preferred code diversity shuffler that may also be used for shuffler  506  in FIG. 32. A timeslot status table in memory  718  stores a current map shared by all RUs and the CU of which timeslots/channels are currently active. In the preferred embodiment, the data stored in this table for each timeslot includes its present mode, its next mode and local/remote information. Permissible modes include: idle where no code is assigned, normal where a code is assigned, fallback # 1  where more than one code is assigned to a timeslot and fallback # 2  where even more codes are assigned to an active timeslot than in fallback # 1  mode. The addresses in table  718  are sequentially scanned using addresses generated on a bus  722  by a counter  720  driven by the chip clock on bus  603 . The data regarding the status of each sequentially scanned timeslot is output on bus  724  to control logic  726 . The status data on bus  724  tells the control logic whether or not a CDMA code needs to be assigned. If control logic  726  sees data indicating a timeslot is active on bus  724 , it generates a signal on bus  728  causing counter/random number generator  730  to generate a pseudorandom number on bus  734  to act as a write pointer for purposes of guiding the encoded 4-bit group from encoder  526  in FIG. 32 into the storage location in buffer memory  548  which will be multiplied by the code pointed to by the number on bus  734 . The code number on bus  734  is generated from a seed number on bus  732 . All RU and CU code diversity shufflers receive this same seed and all RUs having active timeslots and the CU operate synchronously to assign the same CDMA code to the active timeslots so that the CU can recover the CDMA spread data transmitted by the RU using the same CDMA code(s) that were used to spread it. The pseudorandom number generated in this manner is output on bus  734  as an address into a code status table stored in random access memory  736 , and is also stored in FIFO memory  742  for later output as a writer pointer on bus  533 . The code status table stores information shared by all RUs and CUs regarding which codes are eligible for use. Some codes may be block from usage because they either do not have sufficient noise immunity or for some other reason are not to be used. The data regarding whether use of the code pointed to by the address on bus  734  is permissible is output to the control logic via bus  738 . If the data on bus  738  indicates the code pointed to by the address on bus  734  is permissible for use, the control logic generates a signal on bus  740  telling counter  720  that it should now generate an address to read the contents of the next address in sequence in the timeslot status table. All active timeslots are assigned a code once per symbol. 
     It is important in the embodiment of FIG. 41 that the contents of the timeslot status table and the code status table be constantly updated by all the RUs and CU so that they all share the same information. Updates of code status and timeslot status are broadcast by the CU on a broadcast channel using message protocol with CRC and ECC bits appended. The messages about timeslot status are stored in event queue  744  which also receives the address pointer on bus  722 . As the address of each timeslot appears on bus  722 , the event queue searches for update messages regarding that timeslot and updates the contents of the timeslot status table via bus  746 . 
     Returning to the consideration of FIG. 32, the buffer memory  548  outputs two information vectors on buses  549   r  and  549   i . The elements in these information vectors are, respectively, the first two bits in every Trellis encoded tribit as the real information vector and the last two bits of every Trellis encoded tribit as the imaginary information vector. 
     In FIG. 32, block  510  generates the ranging Barker codes needed for the ranging process to achieve frame synchronization. In the CU transmitter of FIG. 32, the ranging circuit  510  generates a constant Barker code of 13 bits at level power transmitted during every CU frame gap. In the RU transmitters, the Barker code is transmitted with varying delays and varying power levels per the data on bus  512  until the gap is hit. Preferably, this ranging Barker code generator  510  is a state machine. Rules for creating this state machine in the embodiment represented by FIG. 32 are: any activity in the gap indicated by the ranging status message that does not indicate the RU&#39;s temporary ID indicates a collision; a simple binary stack contention resolution algorithm is used where once an RU starts ranging, any subsequent collision push it deeper on the stack and any empty gap pops it closer to the top of the stack as in a LIFO mechanism. The ranging state machine  510  also receives as its input on bus  512  from CPU  405  a P parameter which sets the power of the ranging pulse and data which defines the Barker code of the ranging pulse. The ranging circuit  510  in the RU transmitter of FIG. 33 will scan all possible Td delays at a first power level which is low in the range of permissible powers and wait for confirmation from the CU that it has hit the gap. If no such message is received, the RU CPU  405  raises the power level to the next level up and scans through all the possible delays again. This process of scanning all possible delays and raising the power to the next level and scanning the delays again is continued until the RU hits the gap. Circuit  510  also receives on bus  512  RU/CU information which tells the circuit  510  whether it is in an RU or CU. The data on line  512  also controls whether a single Barker code is transmitted or a specific sequence of Barker codes during successive gaps to make up the authentication or signature sequence. The data on bus  512  also controls the position of a Barker code pulse relative to the center of the gap. Since this data comes from the CPU  405 , the CPU knows when the transmitter is ranging and can properly interpret ranging status messages broadcast by the CU and received by the CPU via bus  1096  and command, communication and control circuit  860  in FIG.  30 . Circuit  510  carries out the ranging process including contention resolution, pulse position modulation, steering and signature transmission described elsewhere herein in some embodiments, and in other embodiments, these processes are carried out by the CPU  405  and circuit  510  in cooperation with each other. 
     In some embodiments, circuit  510  in FIG. 32 also plays a role in the upstream equalization process. Upstream equalization is the process of reducing or diminishing undesired noise in the desired upstream data caused by, for example, reflections from impedance discontinuities in the coax or other media, misalignment of frames etc. Equalization is implemented in part by circuit  510  in placing a particular, predetermined pattern of signals in one or more gaps between frames so that the CU and RU receivers can determine the noise characteristics then present in the channel and take steps to “equalize” or reduce the noise. In some embodiments, this is done by the RU adjusting coefficients of an adaptive filter so that it has a transfer function which is the inverse of the transfer function of the channel, i.e., the transfer function of the equivalent circuit representing the media connecting each RU to the CU. Performing equalization increases the overall system throughput capacity, but it is not absolutely essential if lower capacity can be tolerated. 
     Block  514  on the left side of FIG. 32 is a register or memory storing command and control data such as the pilot channel signal to be transmitted on the 16 access and command and control channels. This data arrives on bus  399  the CPU  405 . Block  516  is a multiplexer which selects between the payload data for the 128 payload channels from the framer  508  on bus  518  or command and control data on bus  520 . Switching between these data streams is under control of timing logic which is not shown. The selected data stream is then output on bus  522 . Typical command and control data includes data messages exchanged between the RU and CU and CU regarding ranging such as “I want to start ranging”, “I found more than one Barker code in the gap, please perform your contention resolution procedure” etc. some of which are described in more detail in the discussion of ranging and contention resolution. 
     Because the 4th bit to be added to each tribit depends upon the state of the tribit from this channel during the last symbol, a memory  528  is used to keep a record of the state of each channel&#39;s 4 bit chip state during the last symbol transmission. This information is supplied to the convolutional encoder via bus  530  as each channel&#39;s tribit is encoded during each symbol. The mode in which the diversity shuffler  506  operates is controlled by the diversity shuffler by a signal on bus  534 . 
     Media Access Control 
     Block  540  represents circuitry to acquire an access channel and carry out media access control communications to implement ISO MAC layer protocols. Since there are only 4 access channels across which all message traffic requesting channel bandwidth and awarding same pass, contentions will occur when more than one RU simultaneously requests bandwidth on the same access channel. Therefore, access channels are acquired according to the following protocol. Each RU transmitter receives a seed number on bus  550  and pseudorandomly selects which access channel to attempt to use and pseudorandomly selects which 6 symbols of a superframe comprised of 12 symbols to send. The RU then sends an authentication code identifying itself in the form of the unique sequence of 6 of the 12 symbols of a superframe of 4 frames, said unique sequence pseudorandomly selected using the seed. All RUs use the same seed, so the likelihood of more than one picking the same authentication code is small. The 6 symbols sent can contain the RU&#39;s message telling the CU how many channels it needs, or a separate message can be sent after access is achieved. The CU listens on all access channels, and during each superframe determines if more than 6 symbols were sent. If so, the CU broadcasts a message on the control channel indicating there is a contention on a particular access channel. The RUs trying to gain access then do the contention resolution protocol described elsewhere herein used for ranging. If only 6 symbols are detected during the superframe, the CU broadcasts a message on the control channel indicating which 6 symbols were found. The CU can include in the broadcast message code assignments for the requested channels in reservation embodiments or, in another embodiment, can simply transmit updates to the timeslot activity table indicating which timeslots or channels have been awarded to the RU which gained access. The RU that sent these six symbols then knows that it has been awarded access, and updates its timeslot activity table which is maintained in the diversity shuffler  506 . All RUs hear the timeslot activity update broadcast message and similarly update their timeslot activity tables. 
     Once an access channel is acquired, circuit  540  may, in some embodiments, present data on buses  542   r  and  542   i  to multiplexer  544  which comprise access control messages. Multiplexer  544  either selects these media access messages on buses  542   r  and  542   i  or the encoded chips from the convolutional Trellis encoder  526  to the code division multiplexer  527  via buses  546   r  and  546   i  and buffer  548 . The multiplexer  544  is controlled by switching control signals from the CPU  405  to edit the contents of the buffer  548  to overlay the 4-bit groups of the access control symbols with the payload data on bus  507  so that the media access control 4-bit groups go into the right addresses of the buffer  548  so as to get spread by the CDMA codes assigned to the access channels. 
     The media access control messages constitute requests from RUs for bandwidth and awards of specific channels to the RUs by the CU in some embodiments. The awards of specific channels to specific RUs implement a reservations scheme and the awards can take many forms such as broadcasts on the control channel of timeslot activity table update messages or specific messages on the access channels in other embodiments. Also, other media access protocols other than the reservation scheme which are described elsewhere herein are also possible through various protocols some of which may require message traffic on the access channels. In an important alternative embodiment, all the different schemes for allocating channels to specific timeslots may be used or combinations of schemes for various groups of channels may be used. In this embodiment, the type of scheme used is programmable by the user, and in a variation of this embodiment, may be changed by the CU computer based upon traffic conditions and the number of contentions and efficiency considerations. 
     Because a reservation scheme is implemented in the preferred embodiment, no contentions occur on the 140 non media access control payload channels so no contention resolution protocols are carried out for these channels. However, contentions are expected to occur on the 4 access control channels shared between all the RUs so contention resolution will have to be carried out in the manner described elsewhere herein. 
     Spreading of the spectrum of the chips from the convolutional encoder  526  is done by orthogonal code multiplexer  527 . This circuit or software routine performs code division multiplexing or orthogonal encoding of the data on each channel by matrix multiplication. It sets the amplitude of the output chips on buses  558   r  and  558   i  based upon matrix multiplication of the orthogonal codes times the elements of the input information vectors on buses  549   r  and  549   i  from buffer  548 . Each of the information vectors on buses  549   r  and  549   i  is individually spread by the orthogonal code multiplexer to generate individual real or inphase and quadrature or imaginary result vectors  409  and  413  in FIG. 42 on buses  558   r  and  558   i.    
     There is only one orthogonal, cyclic code that has 144 different codes. That code is used and is, in hexadecimal representation: 0218 A503 BA4E 889F 1D92 C1F3 AB29 8DF6 ADEF. Other codes can be used, but the above code is best. Although cyclic codes are the preferred embodiment for ease of implementation, any other orthogonal, noncyclic code set can also be used in alternative embodiments, or other orthogonal, cyclic codes can be used where fewer channels/timeslots are required. The cyclic code given above uses the convention that all logic 0&#39;s represent −1s and all logic 1s represent +1 in the orthogonal code spreading matrix. The first code of the 144 different codes in the code set will be all 1s regardless of the contents of the code given above. The second code in the code set is the code given above: 0218 A503 BA4E 889F 1D92 C1F3 AB29 8DF6 ADEF. The third code is obtained by shifting the code one binary place and taking the overflow bit that “falls off” the most significant bit position edge of the code in the second least significant bit position. The fourth code is obtained by repeating process done to obtain the 3rd code on the 3rd code. 
     The results of the matrix multiplication performed in the orthogonal code multiplexer  527  are coupled via buses  558   r  and  558   i  to one input of a switching circuit  556  switching of which is controlled by the CPU  405 . The other input of the switching circuit  556  is coupled to buses  558   i  and  558   r  to receive the ranging data from ranging circuit  510 . The switch  556  selects the data on buses  558   r  and  558   i  for coupling via buses  557   r  and  557   i , respectively, to a precode FFE/DFE filter  563  during the three symbol transmission times of each frame when payload data is being sent. The switch  556  selects the ranging pulse data on bus  560  during the gap following transmission of the last symbol in each frame. 
     Equalization, as that term is used herein, is the process of compensating for distortions and noise that occur caused by noise in the channel between each RU and the CU. The precode filter  563  performs a measured predistortion at each RU transmitter so that the data arrives at the CU undistorted despite the channel impairments between that particular RU and the CU. The amount of the predistortion is calculated by each RU to substantially or exactly compensate for the current distortion conditions existing in the channel between it and the CU. The predistortion characteristic is implemented by setting the transfer function of the precode equalization filter  563  by changing the tap coefficients of the filter. This transfer function is controlled by the RU/CU Coefficient data input to the filter on bus  561 . Each RU uses its own unique, measured RU/CU Coefficient data to establish a predistortion which is appropriate to its own signals for its position on the network so as to cause its signal to reach the CU with little or no distortion. More details on both upstream and downstream equalization are given in connection with the discussion of the training process symbolized by the flowcharts of FIGS. 45A,  45 B and  45 C.  53 A through  53 C. 
     The CU transmitter of FIG. 32 differs from the RU transmitter in the sense that the precode filter  563  has its tap coefficients set to implement an average predistortion suitable for transmission to all RUs. This predistortion transfer function can be set by averaging the individual predistortions calculated individually for each RU. 
     The output of the precode filter on buses  562   r  and  562   i  is applied to a scaler amplifier  564  which scales the amplitude level of the digital numbers on buses  562   r  and  562   i  in accordance with a signal on bus  566  which indicates the activity level of the modem, i.e., how many timesiots are currently in use by this modem. The purpose of this scaling is to enhance performance by taking advantage of the full precision of a digital to analog converter  576  at the output of the transmitter. A digital to analog (D/A) converter has a dynamic range for its analog output. When few timeslots are active, the summation of the CDMA spreading matrix multiplication partial products does not lead to chip amplitudes which extend to the full limits of the D/A converter&#39;s dynamic range. As a result, the full precision of the D/A converter is not used, and the inherent noise of the D/A conversion process affects the transmitted signal more. To make use of the full precision of the D/A converter, scaler  564  “amplifies” the incoming signal based on the activity level such that the resulting swing in digital values going into the D/A converter  576  causes output analog signals which swing between the limits of the dynamic range of the D/A converter. These signals are later reduced in amplitude by a circuit (not shown) which limits the amplitude swings to prevent interfering with other signals sharing the media. 
     The output of the scaling circuit on buses  568   r  and  568   i  are coupled to shaping filter  570  which doubles to perform carrierless amplitude and phase modulation. There are two filters in the shaping filter which have transfer functions which are the Hilbert transform of each other and which have rolloff characteristics set to digitally filter the data on buses  568   r  and  568   i  to limit the bandwidth of the signal on each bus to the width and center frequency of the 6 mHz channel devoted to digital data communication on the coaxial cable or other media  24 . The shaping filter has a squared raised cosine filter characteristic suitable to shape the outgoing chip pulses so as to satisfy Nyquist criteria in a known manner so as to provide optimal signal-to-noise enhancement and so as to minimize intersymbol interference. The filters in shaping filter/modulator  570  can have other transfer functions also which shape the chips to be transmitted such that the spectrum of the outgoing signals satisfy the Nyquist criteria. Any of these other pulse shapes will suffice to practice the invention. The coefficient data on bus  572  provide ability to set and change the filter characteristics of shaping filter/modulator  570 . More details on the operation of the shaping filter/modulator  570  are given in connection with the discussion of FIGS. 42,  43  and  44 . 
     The output of the filter/modulator is coupled on bus  574  (the filter/modulator  570  sums the orthogonal real and imaginary signals after filtering to generate a single signal on bus  574 ) is coupled to the input of the digital to analog converter  576  for conversion to an analog signal for application to the input of an up/down frequency converter  577 . The purpose of the up/down frequency converter is to convert the frequency of the transmitted signal to the frequency allocated for upstream or downstream transmissions as the case may be in accordance with the frequency plan for the shared transmission media. The up/down converter outputs its signal on the transmission media  412  such as coaxial cable, cellular system, satellite uplink etc. 
     Alternative Ranging, Contention Resolution and Authentication Processes Carried Out by RUs and CU 
     Referring to FIG. 45, there is shown a flow chart for a method of ranging using contention resolution where the span of the system is such that all RUs can align to the same gap at the end of one frame. In the preferred embodiment, the ranging, contention resolution and authentication processes of FIGS. 45-47 are carried out through cooperation of the RU receiver of FIG. 30 including the C 3  circuit  860 , frame detector  882  and the R/Tng circuit  763 , CPU  405  and the CU transmitter of FIG. 32 including Rng circuit  510  and the CU receiver circuit of FIG. 31 cooperating with microprocessor  405  and the frame detector  882 . 
     The starting point of the ranging process is block  600  in the RU ranging process shown in FIG.  45 . Block  600  is reached after an RU has powered up and performed a self test and found itself to be operable. Next, test  602  is performed to listen on the control channel to wait until it is clear for transmission (“E”).  602 . If test  602  determines that a collision (“C”) is occurring on the control channel or a single RU is transmitting (“S”) on the control channel. Test  602  vector processing to block  604  when the control channel is free. Block  604  represents the process carried out by circuit  510  in FIG. 32 of transmission of a ranging pulse (typically a copy of the Barker code transmitted in every frame by the CU). The multiplexer  556  is switched to select input bus  560  before transmission of the ranging pulse. 
     After the ranging pulse is transmitted, the CU receiver listens in the gap to determine if it finds a ranging pulse in the gap, and, if so, if only one ranging pulse is present. Block  604  vectors to test  606  after transmission of the ranging pulse in order to listen on the control channel. The CU will transmit an S on the control channel if a single pulse is found in the gap, and will transmit an E on the control channel if the gap is found to be empty. If test  606  hears an S on the control channel, processing is vectored to block  608  to start the authentication process. If block  606  hears an E on the control channel indicating the gap is empty, processing vectors to block  610  to move the ranging pulse plus  8  chips, and processing vectors back to block  604  to send a new ranging pulse. Processing then vectors back to test  606  to listen on the control channel again. This loop continues until either an S for single pulse is heard on the control channel or a C for collision is heard. The CU sends a C when it hears more than one ranging pulse in the gap. 
     When test  606  hears a C, processing is vectored to block  612  to start the contention resolution process which is then performed as symbolized by block  614 . Contention resolution continues until only one pulse is found in the gap or no pulse is found in the gap. If, as a result of contention resolution, no pulse is found in the gap, the CU sends an E on the control channel, which vectors processing to block  616 . Processing then vectors to block  610  to move the ranging pulse 8 chips forward, and the process repeats itself. 
     An Authentication Process 
     Authentication is started when the CU sends a message on the control channel that it has found a ranging pulse from a single RU in the gap. In both embodiments, the gaps of multiple frames are used to send an authentication code. Each RU that has been attempting to synchronize hears the “S” on the control channel in step  606  in FIG. 45 indicating the CU has detected the ranging pulse from a single RU in the gap, and vectors processing to the authentication process represented by block  608 . There are several possibilities for how authentication is performed. The flow chart of FIG. 46 represents one embodiment which uses pulse position modulation to send the authentication code. In this embodiment, each RU that has been attempting to establish synchronization sends one ranging pulse during the gaps of each of 8 frames but varying the position of the pulse in the gap during each gap. In another embodiment previously described, the RU sends an authentication Barker code sequence comprised of sending the Barker code during some gaps of the 8 frame authentication sequence but not during others in a predetermined sequence. Each RU has a unique sequence, but all RUs send pulses during only half the authentication sequence gaps. 
     A Contention Resolution Process 
     Referring to FIG. 46, there is shown a flow chart of a typical process for authentication by CU modems when one RU&#39;s ranging pulse is found in the gap. The authentication process begins at block  608  and immediately proceeds to block  620 . There, the CU sends out an S on the control channel indicating that it has found a single RUs ranging pulse in the gap. Which RU it is is not clear at this point, and the purpose of the authentication process is to determine which RU has hit the gap and so notify that RU so it can freeze its delay at the delay that hit the gap. Before starting the process of determining the RU identity, the CU sends out a command on the control channel for all RUs who are ranging to move their ranging pulses plus or minus the number of chips separating the ranging pulse the CU saw from the middle of the middle 8 chips of the gap. In block  620 , this process is signified by the phrase “send course alignment data to RU to center ranging pulse”. Because ranging pulses from other RUs may also be in the gap, but at an edge, when they also move the position of their ranging pulses, their pulses may also land somewhere in the middle 8 chips of the gap. Since authentication requires that only one ranging pulse be in the gap, block  620  looks for a so-called “edge pulse” or neighbor in the gap in addition to the single pulse previously found so as to make sure there is truly only one ranging pulse in the gap so as to avoid ambiguity. That is, the CU looks to find out if another RUs pulse which was originally in the gap but outside the middle 8 chips has landed in the middle 8 chips after the position of the pulse which was originally found in the middle 8 chips has been moved to the center of the gap. The CU looks for these extraneous pulses first by commanding a shift in the ranging pulse originally found in the gap which led to the broadcast of the S on the control channel to move sufficiently to land in chip  0  of the middle 8 chips. Then test  622  looks for more than one pulse as described in the next paragraph. Then, the CU commands a move of the original ranging pulse to the other extreme, i.e., to move to chip  7  of the middle 8 chips, and the process of test  622  is repeated. 
     The determination of whether more than one ranging pulse is in the middle 8 chips is performed by test  622  which counts the ranging pulses in the middle 8 chips of the gap and determines their locations. If the count of the number of ranging pulses found in the middle 8 chips is greater than one, the CU broadcasts a C on the control channel indicating a collision state, which causes all RUs to vector processing to their contention resolution protocols, as symbolized by block  624 . If test  622  determines that the pulse count is 0 or their is a position error in the position of the single pulse found in the middle 8 chips, test  626  is performed to determine if the number of retries exceeds the maximum allowable number. If not, the process of block  620  is performed again to send new course alignment data to the RUs on the control channel. If the number of retries found by test  626  is found to exceed the maximum, the process of block  628  is performed where the CU broadcasts an E on the control channel indicating the gap is empty. This causes all RUs trying to synchronize to return to their ranging processes and start over at block  600  in FIG.  45 . 
     Once test  622  determines that there is only a single Rus ranging pulses in the gap and it is within the middle 8 chips, processing is vectored to test  630  which determines if noise has caused detection of what was thought to be a ranging pulse but which was only noise. This test is performed by determining if at least two out of three ranging pulses were received when the ranging pulse was commanded to move to the extreme left edge, the extreme right edge and the center of the middle 8 chips of the gap. If ranging pulses were detected at at least two of these three positions, no false alarm exists, and processing is vectored to block  632 . If a false alarm is detected, processing is vectored back to test  626  to start over in positioning the ranging pulse. 
     The process symbolized by block  632  is the process of the CU broadcasting an A on the control channel which signals all RUs that are attempting to synchronize to send their authentication codes. Therefore block  632  states State=Auth which means that the CU is broadcasting an implicit request for the authentication ID (AUID) of the RU whose pulse is in the gap. In response, all the RUs trying to synchronize send their AUIDs in the form of four ranging pulses during the gaps of each of the next four frames of a superframe, each ranging pulse being located in a specific one of the 8 chips positions of the middle 8 chips in the gap. The positions and sequence during these four gaps of the authentication superframe tell the CU which RU has hit the gap. This is the meaning of the language in block  632  “Look for one pulse in each gap [one SF, Pulse Position Becomes No. 1-7]”. The steps following block  632  just check for errors in this process. Specifically, test  634  is performed after each frame to increment a pulse counter and determine if the pulse count has reached 4 by the end of the superframe. If the pulse count is 4 at the end of the authentication superframe, test  634  vectors processing to block  636  where the CU broadcasts an FAE message on the control channel indicating authentication is finished and sends the AUID code out on the control channel for recognition by the RU that sent it. The AUID will be a sequence of 4 numbers from 0-7 which indicate in which chip of the middle 8 of the gaps of the authentication superframe each ranging pulse was found. Each RU that is attempting to synchronize will compare this sequence of 4 numbers to the 4 numbersof its AUID. If there is a match, that RU will know that it successfully hit the gap and will freeze its transmit delay timing at the number that puts its ranging pulse in the center of the 8 middle chips of the gap. Step  638  is then reached indicating that authentication is complete. 
     If test  634  determines that the pulse count is less than 4 after any gap in the authentication superframe is complete, processing is vectored to test  640  to determine if the number of retries exceeds the maximum allowable number. Test  640  sends processing back to block  632  to look for pulses in the authentication superframe gaps and record their positions until the superframe is over and the pulse count is less than 4. Some number of superframes with the RUs sending their AUIDs can be allowed in some embodiments. Eventually, the number of retries exceeds the maximum, and processing is vectored by test  640  to block  642 . In block  642 , the CU broadcasts an E on the control channel and, in response, all the RUs attempting to synchronize will return to the ranging process. 
     Likewise, if at any time, the count determined by test  634  exceeds 4 during the authentication superframe or at the conclusion thereof, an error has occurred or another RU has moved its ranging pulse into the gap. If this happens, test  644  is performed to determine if the maximum number of retries has been exceeded. If not, processing returns to block  632 . Typically, more than one authentication superframe will be permitted with the RUs sending their AUIDs during each superframe. Eventually, after several superframes, if block  636  is not reached, test  644  will trigger vectoring of processing to block  646  where the CU broadcasts a C on the control channel indicating a collision has occurred thereby causing the RUs to return to their contention resolution protocols. 
     Referring to FIG. 47, the ranging and contention resolution protocol performed or the CU side is detailed in flow chart form. Ranging starts with block  650  where the CU sends out a unique Barker code. This Barker code is a unique pattern of data, which, when received by the RUs is echoed by them back toward the CU after imposing a programmable delay. It is this programmable delay that is being adjusted during the ranging process until the echoed Barker code in the form of a ranging “pulse” hits the gap. Block  652  represents the process carried out by the CU of monitoring the gap to determine if any RUs ranging pulse has hit it. This monitoring is typically done by performing a correlation calculation between any signal received in the gap and the Barker code originally transmitted, but in other embodiments, it can be any other form of monitoring such as threshold comparison etc. which is effective given the noisy environment. Threshold monitoring of sharp or high power pulses is less desirable however, because sharp pulses tend to splatter the band with a broad range of Fourier components, while high power ranging pulses that will rise above the noise can, before alignment is achieved, arrive on top of or with payload data from other RUs and interfere therewith. Test  654  represents the examination of the results of the correlation calculation or other monitoring activity to determine if any pulse was found in the gap. If not, step  656  is performed where the CU broadcasts an E on the control channel indicating the gap is empty, thereby causing the RUs to adjust their delays and resend their Barker codes or ranging pulses during the next frame. Step  656  also subtracts one from an iteration stack which counts the number of iterations or attempts to range. Then the monitoring step  652  is performed again. 
     If test  654  determines that there is a ranging pulse in the gap, processing vectors to test  658  where the CU determines if there is more than one ranging pulse in the gap. If there is only one ranging pulse in the gap, step  660  is performed where the CU broadcasts an S on the control channel indicating to all RUs that are ranging to begin their authentication processes. 
     If more than one ranging pulse is found in the gap, step  662  is performed to broadcast a C on the control channel indicating to the RUs that there is a contention and forcing them to carry out their contention resolution protocols. The CU then checks the status of an iteration stack to see if it is full. The iteration stack is used to keep track of the rounds of ranging for purposes of contention resolution and more rapid ranging of all RUs attempting to synchronize in some embodiments. The stack is incremented by one, and tested in test  664  to determine if the maximum number of iterations has been reached. If not, processing returns to block  652  to again monitor the gap for ranging pulses transmitted during the next frame. If the maximum number of iterations has been reached, step  666  is performed to broadcast an R on the control channel thereby causing all RUs to reset and start the ranging process again. 
     Referring to FIG. 49, there is shown a flow chart for a ranging process carried out by the RUs using a binary tree algorithm. The process starts with one or more RUs that are not in frame synchronization but which wish to achieve frame synchronization so as to be able to send data to the CU. These RUs first must synchronize their receivers to broadcasts on the control channel from the CU so that they can receive status commands from the CU which control their activities during the ranging process. The RUs can synchronize to the CU broadcasts themselves without assistance from or the need to send anything to the CU by recovering the system clock signal from the periodic broadcasts of the Barker code signals every frame from the CU. Once this has happened, test  668  determines that control channel signals can be received and ranging can start. Until this happens, path  670  is taken to wait state  672  and block  674  to idle until the RU receiver synchronizes to the CU and can receive its broadcasts. 
     When RU receiver synchronization has been achieved, step  676  is performed to pick some arbitrary delay and send a ranging pulse using that delay. Test  678  is then performed to switch on the control channel signal and determine the state of the CU. If the CU did not find any ranging pulse in the gap, it broadcasts an E on the control channel. Each RU then changes its delay by adding 8 chip times, as symbolized by block  680 , and transitions to step  676  to send another ranging pulse. This process continues until one or more RUs set their delays such that their ranging pulses arrive in the gap. If the CU detects a single pulse in the gap, it broadcasts an S on the control channel which the RUs interpret as an authentication command. Each RU then transitions to step  682  to begin the authentication process, which has been previously described. Basically, the authentication process involves the RU sending its identification code as either a unique sequence of ranging pulse positions in the middle 8 chips of the gaps of multiple frames or as a unique sequence of the presence and absence of ranging pulses in the gaps of multiple frames. 
     If multiple RUs hit the same gap, test  678  finds that the CU is broadcasting a C on the control channel indicating that the RUs need to perform their contention resolution protocols, as symbolized by block  684 . As symbolized by test  686 , each RU then “flips a coin” to determine if it should continue and examines the outcome. If an RU decides not to continue, processing in that RU transitions to test  688  where the RU determines the control channel signal type. If an E is being broadcast, it means that all RUs that were ranging decided to stop, and processing returns to step  686  to “flip the coin” again. If test  688  determines that any other signal is being received, processing returns to block  672  and the ranging process starts over for that RU. 
     If the coin toss results in the RU deciding to continue ranging, step  690  is performed to send another ranging pulse. Then test  692  is performed to listen to the control channel and determine what the CU state is. If the CU found no pulse in the gap, step  694  is performed to move the ranging pulse, i.e., adjust the transmit frame timing delay, and try again. Accordingly, processing transitions back to test  668  through step  672 . If the CU is broadcasting a C, more than one pulse has been detected in the gap, and processing returns to step  686  to flip the coin again to decide whether to continue ranging. If test  692  determines that the CU is broadcasting the S or authentication command, processing transitions to step  682  to begin authentication. After authentication, the CU sends fine tuning commands over the control channel to the RU which just authenticated itself to adjust the position of its ranging pulse to the center of the gap. 
     Referring to FIG. 48, there is shown a flow chart of the preferred process of ranging and contention resolution in the RU using a binary stack. This process is slightly faster than the binary tree algorithm in achieving alignment because in this process, the RU remembers upon which iteration it “failed”, i.e., the coin toss after a contention caused the RU to stop attempting ranging. The process starts with step  698  to listen on the control channel. When a C is broadcast by the CU, step  700  is performed to initialize a binary stack to 0. This stack is used to keep track of the iteration number when the coin toss resulted in a decision to discontinue ranging. Next, step  702  “flips the coin” to make the decision as to whether to continue. If the decision is to not continue, step  704  is performed to push down the stack by setting the value on the stack to stack +1. Then test  706  is performed to listen again on the control channel and determine the CU state. If there is still a contention, step  704  is performed to increment the stack again. If test  706  determines that the CU says the gap is empty or only a single ranging pulse is in the gap, step  708  is performed to pop the stack, i.e., to set the stack value to stack −1 in step  708 . Next, test  710  is performed to determine if the stack value has reached 0. If it has, processing returns to step  702  to flip the coin again to decide whether to resume ranging. If test  710  determines that the stack has not reached zero, test  706  is performed again to listen on the control channel. 
     Returning to the consideration of step  702 , if the original coin toss caused the RU to decide to continue ranging, step  712  is performed to send a ranging pulse. Then test  714  is performed to listen on the control channel to determine the CU status. If a C is being broadcast, more than one RU is in the gap, and processing returns to step  702  to flip the coin again. If an E is being broadcast, the gap is empty and the delay for the next ranging pulse is adjusted by moving the pulse +8 chips and restarting the ranging process in step  716  by transitioning to step  600  on FIG.  45 . If test  714  determines that the CU is broadcasting an S meaning a single pulse has been found in the gap, processing vectors to step  718  to begin the authentication process. 
     Preferred RU Receiver Block Diagram 
     Referring to FIGS. 30 and 31, there are shown detailed block diagrams of the preferred organization for an SCDMA receiver for the RU and CU modems, respectively. Circuits in the receiver which have similar functions have the same reference numbers and will be discussed without distinguishing whether they perform their function in the RU or CU if their functions are identical. Differences in the circuitry will be individually discussed. 
     The RF signals arrives at the receiver on coaxial cable  412  or other media. An RF demodulator section  750  synchronously demodulates the RF signals in the case of the RU receivers using a detector like that shown in FIG. 29 and a local carrier reference signal which is synchronized in phase and frequency to the master carrier embedded in the pilot channel data from the CU. A separate tracking loop in the RU receiver comprised of slicer  800 , a low pass filter (not shown), control loop  781 , VCXO  808  and frequency synthesizer  760  generate the local carrier signal on line  762  so as to be phase coherent with the master carrier. In the case of a CU receiver, the data from each RU is detected by achieving synchronization with the RU carrier using the preamble data sent in each timeslot by the RU prior to sending payload data and using the rotational amplifier  765  and G 2  amplifier  788  to correct for amplitude and phase errors. The demodulator in RF section  750  of the CU receives a synthesized local carrier signal on line  762 . This local master carrier signal is synthesized by frequency synthesizer  760  from the master carrier signal from the CU transmitter section which arrives on line  187 . 
     The RF demodulator  750  outputs an analog signal on line  752  carrying the chip amplitude information for all time slots. The RF section  750  also includes a passband filter having a center frequency centered on the frequency of the 6 mHz wide band carrying the chip data and having a 6 mHz bandwidth. The RF section also includes a variable gain amplifier that has a gain control input coupled to line  758  coupled to automatic gain control circuit  756 . The AGC circuit works over a fixed interval and counts the number of times the input signal is above a preset threshold and the number of times it is below it. A counter is preset to a negative value at the start of the interval. Each time the threshold is exceeded, the counter in incremented. If the counter has counted up to zero at the end of the interval, the AGC gain is set correctly. Positive values call for decreased gain, and negative values call for increased gain. 
     The signal on line  752  is converted to digital information by A/D converter  754  which performs IF sampling as is known in the prior art was first described by Colinberg, whose papers are hereby incorporated by reference. The sampling rate is 4 times the symbol period. The advantage of using IF sampling is that it allows the use of one A/D converter to sample both the sine and cosine carriers. In alternative embodiments, two A/D converters may be used, each having a sample rate substantially greater than the symbol period. IF sampling is not critical to the invention and other techniques of digitization which are compatible with the system may also be used. 
     The gain of the signal represented by the digital data output by the A/D converter  754  is examined by automatic gain control (AGC)  756 , and if the amplitude is not high enough, the AGC circuit generates a signal on line  758  to increase the gain of the variable gain amplifier in the RF section. 
     Phase separation of the sine and cosine components of the QAM modulated data represented digitally on bus  760  is performed by matched filter  761 . The matched filter has two filters which have filter characteristics that are the mirror image of the squared raised cosine filter characteristics of the filters  1134  and  1136  in the shaping filter/modulator  570  shown in FIG.  42 . The matched filters separate the orthogonal real and imaginary components in the received signals and transmit them to the frame detector via buses  904  and  906  in FIGS. 30 and 31. The filter characteristic of the matched filter is established by data from the CPU  405  on bus  1090 . In the preferred embodiment, the output of the matched filter  762  on bus  840  is filtered by an FFE/DFE filter  764  which functions to cut down on precursor and postcursor intersymbol interference. The FFE/DFE filter  764  has the structure of FIG. 50, and each of the FFE and DFE equalizers is an adaptive FIR filter. Adaptive FIR filters and many of the other digital signal processing components of the circuitry disclosed herein are known and are discussed in detail in Elliott,  Handbook of Digital Signal Processing: Engineering Applications , (Academic Press, Inc. San Diego, 1987), ISBN 0-12-237075-9, which is hereby incorporated by reference. In the preferred embodiment, the FFE filter  764  is placed between circuits  765  and  767  to filter the data on bus  769  and that is the purpose of the notation “FFE” inside rotational amplifier circuit block  765  to symbolize this embodiment. In the RU receivers, the coefficients of the FFE/DFE or individually established by the equalization training process described below. However, in CU receivers, the coefficients of the FFE/DFE filter are established as an average for all RUs. 
     Next, despreading of the data and reassembly of the appropriate data into the corresponding timeslots to undo the code shuffling that happened in the transmitters is performed. The first step in this process is accomplished by CDMA MUX  766 . This multiplexer multiplies the incoming data by the transpose code matrix C T  of the code matrix used by CDMA MUX  527  in the transmitter that sent the data. The resulting despread data is stored in buffer memory  768  sequentially in the order of the individual code multiplications. The CDMA MUX  766  or control logic  1082  generates suitable read/write control signals to cause buffer  768  to sequentially store the despread data on bus  776  output by the CDMA MUX  766 . A deshuffler circuit  770  receives the same seed number on bus  772  as was received by code diversity shufflers  506  in the transmitters. The seed number is sent on the control channel, and is relayed to circuit  770  by the CPU  405 . The deshuffler uses the seed number to generate the same pseudorandom numbers as were generated from this seed during every symbol time by the transmitter. These pseudorandom numbers are used to generate read address pointers on address bus  774  which are coupled to the address port of buffer  768  along with suitable read/write control signals. The data stored at the addresses indicated by the read pointers is then output by the buffer on bus  795 . This bus is coupled to one of two inputs of a switch/multiplexer  791 . Because the address pointers are generated in the same sequence as in the transmitters when shuffling data, the data read out of the buffer  768  is read out in the correct sequence to put the despread data back into the sequential order of the timeslots. 
     Other data received by the code shuffling circuit  770  on bus  772  are the Tss data indicating which timeslots are currently being received, and RI indicating which codes are reserved and cannot be used by this RU or CU. 
     This deshuffling operation is not necessary if the receiver is located in an RU because the CU does not use code hopping for data it sends to the RUs. Therefore, in the preferred embodiment of RU receivers buffer  768  and deshuffler  770  do not exist. These circuits are present in FIG. 30 to symbolize the embodiments wherein code hopping is done by the CU. In the CU receiver, these circuits do exist and the function as described. In some embodiments, these circuits do exist, but are not used and a switch  791  guides the despread data on bus  776  from the CDMA MUX  766  around buffer  768  and directly into the input of the G 2  amplifier  788 . An RU/CU signal on line  793  controls the state of switch  791  such that either the data output bus  795  of buffer  768  or the bus  776  is coupled to input  789  of the amplifier  788 . If the receiver is in a CU, bus  795  is coupled to bus  789 , while if the receiver is in an RU, bus  776  is coupled to bus  789 . 
     In some embodiments, the despread data on bus  776  is simultaneously read by a crosstalk detector which functions to determine the amount of interference between adjacent codes and also plays a role in clock recovery so that all RU and CU receivers and transmitters can be synchronized to the same clock. Crosstalk between channels encoded with adjacent cyclic, orthogonal codes always comes from adjacent channels and happens when the data encoded with adjacent cyclic CDMA codes do not arrive precisely aligned in time. In other words, to have zero crosstalk, the clock time at which the first chip of a symbol transmitted on one channel spread with a cyclic CDMA code arrives at the receiver must be exactly the same time as the clock time at which the first chip of a symbol transmitted on an adjacent channel spread with an adjacent cyclic code. This requires precise frame synchronization to minimize crosstalk between channels. A slippage of one chip clock means complete overlap and total crosstalk since adjacent cyclic codes are generated by shifting the code by one place to the right. A slippage or misalignment of less than one complete chip clock will mean that some crosstalk exists. The crosstalk detector in these alternative embodiments detects the amount of crosstalk affecting each chip of each channel by subtracting the amplitude of the chip of the channel currently being processed from the amplitude of the corresponding chip encoded on the immediately preceding channel. 
     In these alternative embodiments, the amount of crosstalk is sent as a clock tracking error to a control loop logic  781  which outputs a clock phase/frequency correction voltage on line  782  (RU receiver only—the following discussion applies only to the RU receiver clock tracking loop in this alternative embodiment). This signal  782  is coupled to the phase/frequency control input of a voltage controlled crystal oscillator  784  in the RU receiver which generates a chip clock reference signal on line  786 . This chip clock reference signal is fed to one input of a switch  787 , the other input of which is coupled to receive an external clock reference signal at 8.192 Mhz. A switching control signal on line  791  from the CPU  405  controls whether switch  787  selects which of the chip clock reference signals on lines  786  or  789  for output on bus  793  to the time base circuit  886 . In some embodiments, the clock signal is multiplied in a PLL (not shown) by a factor of two so that two clock signals can be fed to the time base circuit  886 . This PLL multiplies the clock reference signal on line  793  to generate two output signals at  114 . 688  Mhz and 57.344 Mhz which are supplied on bus  888  to a time base generator  886 . The time base generator generates the various clock signals needed for synchronization of the system, and these clock signals are coupled to every circuit in the receiver and transmitter which need them. 
     In the preferred embodiment for an RU receiver however, clock recovery is performed in the RUs by frame detector  882  using the fine tuning circuitry shown in FIG.  34 . This circuitry generates a clock steering tracking error signal on line  900  in FIG.  30 . This clock steering signal is input to the digital equivalent of an integrator in control loop  781  which serves as a loop filter for a phase lock loop including VCXO  784 . The averaging process of integration eliminates the random noise. The integrated error signal is output as a clock phase steering signal on line  782  to the error signal input of VCXO  784  to generate the clock reference signal on line  786 . The CU receiver of FIG. 31 does not have a clock tracking loop like that just described. 
     Although a global automatic gain control adjustment was made by AGC  756 , data is being received in the CU receiver from many different RUs located at many different positions on the network. To minimize errors in interpretation of the upstream received data caused by amplitude variance caused by differing path length losses from the various RUs and channel impairments, a separate gain control adjustment is desirable for each RU. This is done by transmitting from each RU a preamble of known data before the payload data for each timeslot assigned to that particular RU as mentioned above. A variable gain G 2  amplifier  788  is employed in the CU to amplify each timeslot&#39;s data with an individual gain value established to overcome the near-far problem so that the data from all the RUs, regardless of their distance from the CU, have the same amplitude level at the slicer  800 . The same G 2  amplifier is employed in the RU, but the gain value is fixed at one value for all the timeslots from the CU so that the CU signals to be adequately strong to be detected in the slicer and Viterbi decoder. Thus, in the RU receiver of FIG. 30, bus  792  is not present since the gain adjustment factor is the same for all timeslots. The control loop logic  781  assists in gain adjustment process in the RUs by sending a desired gain signal on line  790  to amplifier  788 . The details of the design of the control loop circuitry  781  are not critical to the invention and any person skilled in the art can design suitable circuitry to function in the manner specified herein for the various embodiments. In the CU receiver, the gain adjustment factor on bus  790  results from the inputs received on buses  792  and  794 . The input on bus  792  in the CU receiver tells the control loop which particular timeslot&#39;s data is currently at the input  789  of the amplifier  788  and is generated by deshuffling circuit  770 . The control loop  781  in the CU receiver of FIG. 31 also receives an input on bus  1086  from control logic  1082  and CPU  405  which indicates when preamble data for a particular timeslot is being received. The input to the control loop  781  on bus  794  is the gain adjustment factor to use and this factor is received by the control loop in both RU and CU receivers. The gain adjustment factor is generated by a memory  796  which stores individual gain control and phase error correction numbers for each of the  128  payload channels (or all 144 channels in some embodiments) in the CU receiver. In the RU receiver, memory  796  stores only one gain adjustment value. 
     During reception of preamble data, the control loop  781  cooperates with the slicer  800 , the G 2  amplifier  788  and the rotational amplifier  765  to carry out an iterative process to reduce the slicer error to as low a value as possible by adjusting the amplitude error and phase error coefficients in the Upstream Carrier Recovery Error Correction Factor equation (Equation (5)) given above. Specifically, the CPU  405  and control logic  1082  will signal the control loop  781  and slicer  800  when preamble data is being received. Notification to the slicer  800  in FIG.  31  and slicer/detector  467  in FIG. 28 takes the form of activation of the CU Preamble signal on line  1086 . When preamble data is being received, the control loop will set initial values for the 1/a and e −jø  amplitude and phase error correction factors of Equation (5) and transmit these on buses  790  and  802 , respectively, to the G 2  amplifier  788  and rotational amplifier  765 . In the preferred embodiment, the G 2  and rotational amplifiers are the same amplifier, but they are shown separately in the figures for clarity of illustration of the concept. These circuits in the CU receiver will operate on the received data samples to make amplitude and phase error corrections. In the CU receiver only, the slicer will compare the received preamble data signal to the 3-j constellation point it knows it is supposed to be receiving during the preamble to derive amplitude and phase correction factors for the particular RU that sent the preamble data. The amplitude and phase errors between the actual received data and the 3-j point are output on bus  798  to the control loop  781 . The control loop  781  examines these error values, and adjusts the 1/a and e −jø  amplitude and phase error correction factors in an appropriate direction to tend to minimize the slicer error. The process repeats itself for the next preamble 3-j constellation point. Eventually, the control loop finds values for the 1/a and e −jø  amplitude and phase error correction factors that minimize the amplitude and phase error values on bus  798 . These values are then recorded in memory  796  in FIGS. 31 and 28 as the 1/a and e −jø  amplitude and phase error correction factors to use in receiving in the CU data for the timeslot(s) assigned to the particular RU for which the correction factors were calculated. The process is repeated for each RU and each time the RU transitions from an idle state to an active state. This process resynchronizes the CU receiver detection process for each RUs data occasionally or periodically without the use of tracking loops in the CU. These correction factors are used only for controlling the G 2  amplifier and rotational amplifier in the CU receiver and are not used to steer any clock or carrier VCXOs in tracking loops. 
     The process described above regarding synchronization in the upstream to the preamble data gives upstream carrier recovery synchronization. Frame synchronization and chip clock synchronization are done in the CU for the upstream data by the frame detector  882  using the coarse and fine tuning circuitry of FIG.  34 . The CU receiver knows when the gap is, so the frame detector  882  in the CU does chip clock synchronization only based upon the RU&#39;s Barker code transmissions. The frame detector  882  in the CU receiver of FIG. 31 also functions to look for ranging Barker codes and supports the process of instructing the RUs on how to alter their transmit frame timing delay values T d  so that their Barker codes hit the gap. 
     After synchronization to the preamble in the upstream data, the CU receiver control loop  781  uses the information received on bus  792  regarding which timeslot&#39;s data is currently being received to generate an address pointer to that timeslot&#39;s amplitude (1/a) and phase error (e −jø ) correction coefficients in memory  796 . The control loop  781  then sends the address pointer to memory  796  via bidirectional bus  794  along with suitable read/write control signals and receives from the memory the amplitude and phase error correction coefficients for the particular timeslot being received. The control loop then places the amplitude and phase error correction coefficients on buses  790  and  802 , respectively, to control the digital amplification process carried out by the amplifier  788  and the phase error correction process carried out by the rotational amplifier  765 . 
     The slicer  800  is of conventional design, and includes circuitry to measure both gain and phase error for each channel&#39;s data. These errors are measured by circuitry in the slicer which compares the amplitude and phase of a received chip to the amplitude and phase of the legitimate constellation point which the received chip is supposed to represent. Recall that the constellation of FIG. 18 represents all the permissible 4 bit chips that can be part of a symbol. Each chip is comprised of 2 bits plus a sign bit which define the real or I axis coordinate and 2 bits plus a sign bit which define the imaginary or quadrature Q axis component. Therefore, in polar coordinates, each constellation point has an amplitude and phase the combination of which defines the constellation point. The circuitry in slicer  800  responsible for quantifying the magnitude and phase errors compares the magnitude and phase of the received point to the most probable point it is supposed to be and generates amplitude error and phase error signals on bus  798  from the differences. 
     The phase rotation amplifier  765  adjusts the amplified data on bus  789  representing each received chip so as to rotate the phase thereof to correct the phase error for that received chip. This is done by a matrix multiplication of the complex number representing each chip by cosine (ø)+j sine (ø) where ø is the amount of desired phase correction. 
     In the RU receiver of the embodiment of FIG. 30, the control loop  781  also uses the phase error data on bus  798  when the pilot channel data is being received to generate a local oscillator steering voltage on line  806  to alter the phase and/or frequency of a 3.584 MHz reference clock output on line  810  by a voltage controlled crystal oscillator  808  (vcxo). The steering signal on line  806  is a carrier tracking error derived from the pilot channel signal. The pilot channel signal carries the master carrier and time synchronization data (such as kiloframe markers) mapped onto a qpsk constellation. The carrier tracking error is extracted based upon a decision directed discriminator. Carrier recovery is started immediately after the AGC gain is set and ranging has achieved frame synchronization. The carrier recovery circuitry just described is monitored by the modem CPU to insure that it remains in synchronization, and if lock is lost, an interrupt occurs which causes re-initialization of the modem to be started and the modem transmitter to be disabled. The same is true if clock synchronization is lost, i.e., the RU local clock is locked to the CU clock and the clock recovery circuitry is monitored to make sure clock synchronization is not lost. 
     In the preferred embodiment, the master clock and master carrier signals generated by the CU modem are generated from the same temperature compensated VCXO by using different multiplication factors to generate the high speed clock and the master carrier signals. Thus, both the master clock and master carrier phase and frequency information are inherently embedded in the Barker codes transmitted by the CU during every CU transmit frame gap. The RUs in the preferred embodiment use the early late gating circuitry in the frame detector  882  in FIG. 30 to recover the master clock signal via the clock steering signal on line  900  and apply this master clock signal to frequency synthesizer  760  to generate the master carrier signal. The recovered master clock signal is supplied to the RU transmitter on line  901  and the recovered, synthesized master carrier signal is supplied to the RU transmitter on line  903 . 
     Once carrier recovery has been achieved, the kiloframe data encoded in the pilot channel is recovered to achieve kiloframe synchronization so that the RU modem registers and software can be initialized to beginning counting CU frames so as to be able to keep straight which assigned codes from CU messages are to be used during which transmitted RU frames. The RU receiver decodes the synchronization sequence data on the pilot channel using a bpsk constellation. The CU transmitter generates the pilot channel signal as pseudorandom synchronization sequence of bits which are taken one at a time, bpsk modulated and transmitted on channel  1 , one bit per symbol or 3 bits per frame. The RU generates its own matching pseudorandom sequence locally in a manner to be described below. The RU receiver frame detector demodulates and decodes the incoming pilot channel bits using its own internal slicer and compares them to its own matching pseudorandom pilot channel sequence. Each bit has only 2 possible digital values which defines 2 points in the bpsk constellation. If the incoming points are rotated in phase from one of these 2 points, the rotation is a carrier phase error and is used to generate a carrier phase steering signal on line  900  (line  900  carries both carrier steering signals and clock steering signals from the early-late gate sampling circuit in the frame detector) in FIG.  30 . The carrier steering signals are transmitted to the control loop  781  and vcxo  808  via bus  806  to keep the local carrier reference signal on line  810  synchronized to the pilot channel data. 
     A kiloframe is 1024 frames long. As the pilot channel bits are received, they are compared to the local pseudorandomly generated pilot channel sequence. If communications were perfect, and frame synchronization is perfect, the incoming bits of the synchronization sequence would match the locally generated sequence exactly. A state machine counts the number of mismatches, and, if it is less than a threshold, frame synchronization is assumed, and the errors attributed to noise on the channel. If the number of errors exceeds the threshold, an interrupt signalling loss of frame synchronization is generated, and re-initialization is started. The kiloframe marker is detected in the synchronization sequence when a 16 bit feedback shift register which is loaded with 16 bits of the incoming bit stream of the synchronization sequence reaches a state which it only reaches after 1024 frames of bits of the synchronization sequence have arrived. 
     Specifically, referring to FIG. 51, the circuitry of the frame detector  882  which monitors frame synchronization in the RU receiver and detects the kiloframe marker in the pilot channel synchronization sequence is shown. The bpsk pilot channel data enters on line  906  and is detected in a slicer  1320 . The slicer output is coupled to a first input of an exclusive-OR (xor) gate which inverts the data selectively to correct phase ambiguity (the carrier may accidently lock in 180 degrees out of phase which causes every bit in the locally generated pseudorandom sequence to be the opposite of the incoming sequence bit) in accordance with a ambiguity signal on line  1324  from state machine  1326 . A switch  1326  under control of the state machine selects the data on line  1328  for input to the 16 bit feedback shift register (FSR)  1330  for the first 16 clock cycles. The FSR is clocked once per symbol by a clock signal on line  1332  from time base  886  in FIG.  46 . After the first 16 incoming bits are loaded, the multiplexer is switched by the state machine to select the feedback data on line  1334  from the output of xor gate  1336  which has its inputs coupled to the two MSB outputs of the FSR. The FSR acts as the local pseudorandom number generator to generate a sequence of bits that is supposed to match the incoming synchronization sequence. The feedback data on line  1334  serves as a predictor of the next incoming bits in the sequence, and is fed to one input of an xor gate  1338 . The other input of this gate receives the actual incoming bits of the synchronization sequence. The feedback bits are also fed back into the FSR through switch  1326  to further alter the state thereof. The xor gate  1338  outputs a zero on line  1340  when the predicted bit on line  1334  matches the actual incoming bit. The zero on line  1340  does not enable error counter  1342 , so no error count incrementation occurs. If the predicted bit disagrees with the actual bit, the error counter  1342  is enabled and incremented on the next symbol clock. Timer  1344  controls the interval over which the error count is taken. The error count on bus  1344  is read by state machine  1326  and used to detect loss of frame synchronization and automatically signal this event and attempt to reacquire the pilot channel and frame synchronization. Re-initialization is initiated by the software upon receipt of a loss of frame synchronization signal from the state machine. Reacquisition is continually retried until kiloframe synchronization is again achieved. 
     FIG. 52 is a state diagram for the state machine  1326  that monitors frame synchronization. The state machine starts in acquisition true state  1352  by controlling switch  1326  to allow sixteen synchronization sequence bits enter the FSR  1330  without inverting them via the ambiguity signal on line  1324 . Transition to qualification true state  1354  then occurs where the error count on line  1344  is monitored and switch  1326  is controlled to select the feedback bits on line  1334  for input to the FSR  1330 . State  1354  determines if for each count interval, the error count exceeds or is less than threshold  1 . If the count exceeds threshold  1 , the possible problem is that the carrier has locked on 180 degrees out of phase. Transition to acquisition invert state  1356  then occurs where the ambiguity signal is driven so as to invert the next 16 incoming pilot channel bits, and switch  1326  is controlled to load these inverted bits into the FSR. Next, the state machine transitions to qualification invert state  1358  where the incoming pilot channel bits are inverted and switch  1326  is switched to select the feedback bits on line  1334 , and the error count is again monitored. If the error count exceeds threshold  1  again, the problem is not a phase ambiguity, so path  1360  is taken to state  1352  to start over and the ambiguity signal is set to not invert the incoming bits. If the error count is less than threshold  1 , the phase ambiguity was the problem, and path  1362  is taken to tracking invert state  1364 . The state machine stays in state  1364  with incoming pilot channel bits being inverted and compared to the predicted bits generated by the FSR as long as the error count remains below a second threshold. As soon as the error rate exceeds threshold  2 , transition to acquisition invert state  1356  occurs and a new 16 inverted pilot channel synchronization sequence bits are a loaded through switch  1326  into the FSR, and the process starts over. A tracking true state  1366  works the same way as state  1364  except where lock on was reached without inverting the incoming pilot channel bits. 
     After  1024  frames of the foregoing activity, the state of the output lines  1346  will be all 1&#39;s. This is the kiloframe marker. This state causes AND gate  1348  to sent a K_F kiloframe marker  1  to training generator  1352 . This circuit is used to coordinate frame tag number counting in the RU receiver. The RU counts incoming CU frames by virtue of a counter which counts the occurrences of the GAP_a signal from the CPU  405 . In the preferred embodiment, this counter is located in the time base  886  in FIGS. 30 and 8 which communicates with the CPU via bus  1350 . The frame counter can also be in frame detector  882  in FIG.  30  and frame detector  513  in FIG. 8 which communicate with the CPU via buses  755  and  902 , respectively. 
     Returning to the discussion of FIG. 30, the carrier reference frequency on line  810  generated from the preamble data is used by frequency synthesizer  760  to generate a local carrier signal on line  762  that match the frequency and phase of the carrier signals used in the QAM modulators in the RU transmitters. Line  762  is coupled to the local carrier input of a synchronous demodulator in RF section  750 . The control loop  781 , VCXO  808  and the frequency synthesizer  760  combine in the embodiment of FIG. 30 to perform the function of the carrier recovery circuit  515  in FIG.  8 . 
     The receiver of FIG. 30 uses two feed forward equalizers (FFE) and two decision feedback equalizers (DFE). The first FFE and DFE are shown combined as circuit  764  designated CE just after the matched filter  761  and just before the orthogonal code demultiplexer. The second FFE is combined with a rotational amplifier in circuit  765  after the orthogonal code demultiplexing operation and before the slicer. The second DFE is circuit  820 . The circuits  820 ,  830 ,  832 ,  800 ,  767  and the FFE portion of circuit  765  are collectively referred to as the SE circuit in the materials included below on power alignment and other issues. 
     The equalization process involves some interplay between these FFEs and DFE as will be described below in the section on equalization. Both of the FFEs function to eliminate or substantially reduce precursor intersymbol interference, and both DFEs function to reduce or eliminate post cursor intersymbol interference. 
     Precursor and postcursor ISI can be understood as follows. If a transmitter were to send an impulse signal on one symbol with adjacent symbols empty, the receivers in an ideal system would receive the impulse with zeroes on either side of it. However, because of channel impairments, the receivers will receive an impulse and there will be some nonzero data in symbols on either side of the impulse. The nonzero data in symbols that precede the impulse symbol in time are precursor intersymbol interference. The FFE circuits removes this interference. The nonzero data in symbols that follow the impulse symbol in time is postcursor interference which is removed by the DFE circuits. The DFE circuit  820  receives as one of its inputs the decision data output by slicer  800  on bus  836  and processes these signals in accordance with the filter transfer function established by the tap weight coefficients received on bus  842  from a least means square calculation circuit. The resulting signals are output on bus  846  to the subtraction input of difference calculation circuit  767 . The DFE and difference calculation circuit combine to subtract out that portion of the intersymbol interference produced by previously detected symbols from the estimates of future samples. 
     All the DFE and FFE circuits are FIR filters with adaptive tap coefficients. There is one main tap designated tap  3  and three secondary taps designated taps  0  through  2 . The DFE circuit  820  and the FFE circuit  765  (circuit  765  is an FFE only during the equalization training period and is a rotational amplifier during payload data reception after training) receive their adaptive tap coefficients on buses  842  and  838 , respectively, from the least mean square calculation circuit  830 . The FFE/DFE circuit  764  receives its tap coefficients via bus  844  from the least mean square calculation circuit  830 . The FFE and DFE FIR filters are given initial values for their adaptive tap coefficients that are close enough to allow the adaptation process to proceed. These preset coefficients are supplied from the CPU  405  via buses  824 ,  821  and  822 . Thereafter, the coefficients are adaptively altered by signals on buses  842 ,  838  and  844  by the least mean squared circuit  830  using a conventional precursor and post cursor ISI elimination tap coefficient calculation algorithm. 
     The least mean square (LMS) circuit  830  iteratively calculates the new tap coefficients in a conventional manner and interacts with the FFEs and DFEs in the manner described below in the equalization section. The LMS starts with the initial tap weights and iteratively calculates the convolution sum between the tap input signals (input signals to each stage of the tapped delay lines) within the FFE  765  and the DFE  820  and the tap coefficients of the FFE  765  and DFE  820 , all of which are obtained via bidirectional buses  842  and  838 . The LMS then receives error signals on bus  831  calculated by difference calculation circuit  832  defined as the differences between the desired data points on bus  836  and the received data points on bus  834 . The LMS then calculates new tap weights by multiplying the error signals times the corresponding tap input signals used to calculate the convolution sum times a predetermined step size which sets the rate of convergence to a stable value, and the result is added to the old tap weights to arrive at the new tap weights. These new tap weights are then sent to the FFE  765  and DFE  820  for use during the next iteration. 
     The LMS circuit implements a calculation which is based upon the fact that the needed change in the adaptive coefficients to the adaptive FIR filters  764  and  820  is proportional to the error on bus  831  times the conjugate of the data being input to the filters. In other words, the error is multiplied by complex numbers representing the received chips which have had the signs of their Q or imaginary components inverted. 
     The DFE filter eliminates or reduces post cursor interference by supplying a subtraction value on bus  846  to subtractor  767 . The data sent by the DFE filter on bus  846  is subtracted from the data on bus  769  output by the FFE filter  765  during the equalization training interval. Eliminating the precursor interference and post cursor interference from the data on the bus  834  allows the slicer  800  and a Viterbi Decoder  850  to make better decisions about what chips were actually sent despite the channel impairments. The LMS, DFE and FFE circuits can be eliminated in some simple embodiments with, for example, only 4 points in their constellations. But to get more data throughput, more complex constellations are needed, and in such a situation, the points are closer together and ISI interference makes decisional discrimination between the constellation points more difficult. This creates a need for the above described ISI elimination circuitry. 
     After correction for ISI interference, the corrected data is passed via bus  834  to slicer  800 . The purpose of the slicer is to make instantaneous decisions regarding which point in the constellation each chip represents for purposes of generating the gain and phase errors needed by the control loop and for purposes of generating the desired data on bus  836 . The slicer does not make use of the 4th redundant bit in each chip for this purpose, and, as a result, makes errors in interpreting chips. It is up to the Viterbi Decoder  850  to correct these errors of interpretation. 
     Viterbi Decoders are well known in the art, and any Viterbi decoder algorithm will suffice for purposes of practicing the invention. The particular Viterbi algorithm used in the preferred embodiment is given below. Basically, Viterbi Decoder  850  and memory  852  keep track of the present and last state for each timeslot for purposes of tracing a path through a three dimensional space defined by the constellation of permissible input points stretched out over a third axis representing time which is orthogonal to the I and Q axes. There is one of these three dimensional spaces for each timeslot. By making use of the redundant bit or bits in each chip, and examining the path the states of each timeslot take through the appropriate 3-D space over time, the Viterbi Decoder makes a better informed decision as to which legitimate point in the constellation of permissible points each received code represents. The information on bus  792  to the Viterbi Decoder from the deshuffler tells the Viterbi Decoder which timeslot during which each code received on bus  836  was transmitted. The Viterbi Decoder uses this information to generate an address pointer to memory  852  pointing to the state information for that timeslot. This allows memory  852  to output the state information which is used by the Viterbi Decoder to make its analysis. 
     In the preferred embodiment, the following Viterbi algorithm is used. 
     
       
         
           
               
               
             
               
                   
               
             
            
               
                 N = 16; 
                 %  number of states 
               
               
                 C = 2; (C=2+3;) 
                  % for trace back in one symbol time. 
               
               
                   
                  % (if trace back is 1/3 in a symbol time C=2+3) 
               
               
                 Dd = 12 
                 % Decision Delay 
               
            
           
           
               
            
               
                 for every input_symbol 
               
            
           
           
               
               
            
               
                   
                 for present_state = 0:N−1 
               
            
           
           
               
               
               
            
               
                   
                 for i = 0:7 
                 % loop on previous states 
               
            
           
           
               
               
            
               
                   
                  previous_state = f(present_state, i); 
               
               
                   
                  tx = f(previous_state,present_state); % possible transmitted signal 
               
            
           
           
               
               
               
            
               
                   
                 bm = f(tx,r); 
                 % branch metric 
               
            
           
           
               
               
            
               
                   
                  pm_tmp(i) = pm(previous_state) + bm; % ADD, find path metric 
               
            
           
           
               
               
            
               
                   
                 % ADD with limiter (no overflow) 
               
            
           
           
               
               
            
               
                   
                 % pm_new(present_state) can be computed here by minimum 
               
            
           
           
               
            
               
                 of 2 values 
               
            
           
           
               
               
            
               
                   
                 % previous_state_min can be computed here with pm_new(present_state) 
               
            
           
           
               
               
            
               
                   
                 end 
               
            
           
           
               
               
               
            
               
                   
                 [pm_new(present_state),previous_state_min] = min(pm_tmp); 
                 % Compare &amp; 
               
            
           
           
               
               
            
               
                   
                 % Select (find min &amp; index), (can be computed in loop of i) 
               
            
           
           
               
               
            
               
                   
                 survivor(present_state,survivor_pointer) = previous_state_min; % update survivor, 
               
            
           
           
               
               
            
               
                   
                  % i (3 bits) can be saved instead of previous_state_min (4 bits). 
               
            
           
           
               
            
               
                  end 
               
            
           
           
               
               
            
               
                  pm = pm_new; 
                  % update path metric, or switch path metric memory 
               
            
           
           
               
               
            
               
                  if mod(symbol,3) = 0, do: 
                 % begin trace back when 1&#39;st symbol of frame is received. 
               
            
           
           
               
               
            
               
                   
                 surv_rd_add_l = min(pm)  ; % survivor RD address 
               
               
                   
                  surv_rd_add_h = survivor_pointer; 
               
            
           
           
               
               
               
            
               
                   
                 start TRACE_BACK; 
                 % While trace back is employed continue the process. 
               
            
           
           
               
               
            
               
                   
                 end 
               
            
           
           
               
               
               
            
               
                   
                 inc(survivor_pointer); 
                 % circular increment survivor_pointer 
               
            
           
           
               
            
               
                  end 
               
            
           
           
               
               
            
               
                   
                 % TRACE_BACK (Two options: 
               
            
           
           
               
               
            
               
                   
                 % a. Trace back all the survivor memory and output 3 symbols in one symbol time. 
               
               
                   
                 % b. For each symbol trace back 1/3 of the survivor memory. 
               
            
           
           
               
               
            
               
                   
                 % the trace back is employed while the ACS is employed too. 
               
            
           
           
               
               
            
               
                  surv_rd_add = [surv_rd_add_h , surv_rd_add_l]; 
                 % RD ADD of survivor memory 
               
            
           
           
               
               
            
               
                  for k = 1:Dd + 2 
                 % trace back loop, 
               
            
           
           
               
               
               
            
               
                   
                 surv_rd_add_l_old = surv_rd_add_l; 
                 % save old address 
               
            
           
           
               
               
               
            
               
                   
                 surv_rd_add_l = survivor(surv_rd_add); 
                 % read survivor memory 
               
            
           
           
               
               
               
            
               
                   
                  surv_rd_add_h = dec(surv_rd_add_h); 
                 % circular decrement 
               
            
           
           
               
               
               
            
               
                   
                 if k &gt;= Dd, 
                 % Get 3 output symbols 
               
            
           
           
               
               
               
            
               
                   
                 out(0:2) = f(surv_rd_add_l,surv_rd_add_l_old) 
                 % output 3 bit symbols 
               
            
           
           
               
               
            
               
                   
                 end 
               
            
           
           
               
            
               
                 end 
               
               
                   
               
            
           
         
       
     
     The branch metrics are calculated after one symbol in normal mode and after two symbols in fallback mode, and then are stored in memory. The precomputed branch metrics are then used to calculate the path metrics. In fallback mode, the branch metrics of the two symbols are computed by summing the two square distances to each QPSK symbol. The branch metrics of the decoded symbols are summed to obtain one branch metric as in normal mode. 
     The particular Trellis code selected for implementation in the invention is rotational invariant with no parallel paths and 16 states. 
     After the Viterbi Decoder  850  outputs the correct data for each timeslot on bus  854 , deframer  856  reassembles the data into the time division multiplexed timeslots in which these same data originally arrived at the framer circuit of the transmitter for encoding and CDMA spreading. The deframer  856  also descrambles the data to undo the effects of the scrambling carried out by the scrambler  524 . The resulting TDMA stream of 9-bit bytes is output on serial data format bus  858 . Each 9-bit byte in this data stream is comprised of the deshuffled, descrambled three tribits into which it was originally broken in the framer of the transmitter to form the three symbols of the frame during which this 9-bit byte was transmitted. 
     The output bus  854  from the Viterbi Decoder  854  is also coupled to a command and control channel circuit  860  which stores and/or processes codes sent on the command and control channels in the downstream data. Some switching or multiplexing function to select the command and control codes out of the stream of data on bus  854  is provided but is not shown. Codes sent on the access channel in the upstream or downstream data are stored and/or processed by an access channel circuit  862  which receives these codes from the output of the Viterbi Decoder  850  via bus  854 . The command control code data is input to C 3  circuit  860  from the Viterbi Decoder via bus  854 . The CPU  405  accesses the command and control data and access channel communications from the C 3  circuit  860  and the access channel circuit  862  via bus  1096 . The processing of the command and control channel codes and access channel codes may also occur in circuits  860  and  862 , respectively, in alternative embodiments without interaction with the CPU, or the codes may simply be buffered in circuits  860  and  862  until they can be read by a management and control process performed in the CPU  405 . 
     The ranging process in its various embodiments described earlier herein is aided by the R/Tng circuit  763 . This circuit receives an RU/CU signal on line  759  from the CPU  405  which tells the circuit whether it is performing its function in an RU or a CU. In the preferred embodiment, circuit  763  is simply a DMA FIFO which stores status information regarding positioning of the Barker codes in the guardbands during the ranging and initial frame synchronization process. This status information is received from the frame detector  882  via bus  883 . This data is relayed to the CPU  405  via DMA transfers over bus  755  to a memory (not shown) coupled to the CPU  405 . If it is performing its function in an RU, circuit  763  stores status data generated by the frame detector circuitry in implementing any of the functions indicated for any selected one of the embodiments of the RU in the ranging, contention resolution and authentication flow charts. This data may include data as to how many ranging pulses appeared in the gap and data to be sent to the ranging circuit  510  in the transmitter via bus  757  for purposes of setting transmit frame timing delay. These messages to the transmitter on bus  757  include data telling the transmitter ranging circuit  510  when the Barker code or other signal from the CU has been received in each frame thereby establishing the receive frame timing reference, whether to transmit another ranging pulse after contention resolution, and how to adjust the delay factor that establishes the transmit frame timing reference before sending each ranging pulse or Barker code, and, in some embodiments, what Barker code to transmit. 
     In the preferred embodiment, command, communication and control (C 3 ) circuit  860  receives message traffic involved in the ranging, authentication and media access control processes as detailed in the flow charts and transmits this data to CPU  405  via bus  1096 . Such data includes data from the CU indicating when authentication is desired and data regarding when to start sending that particular RUs authentication code. Circuit  860  also receives the authentication code broadcast by the CU after an authentication interval to determine if it is the RU that hit the gap. If so, circuit  860  sends a message to the transmitter via CPU  405  to freeze its current value for the transmit frame timing reference delay at the value last used for transmission of the ranging pulses in the authentication code sequence. The circuit  860  also monitors the control channel for instructions from the CU on how to adjust its transmit frame timing reference delay to exactly center the ranging pulse in the center of the gap. 
     If the signal on line  759  indicates the receiver of FIG. 30 is operating in a CU, the circuit  860  and the CPU  405  carries out those functions indicated for any selected one of the embodiments of the CU in the ranging, contention resolution and authentication flow charts. Circuit  860  and the CPU  405  in the RU and CU combine to process the following data in support of ranging, authentication, contention resolution and fine tuning: data received from the frame detector  882  and R/Tng circuit  763  regarding how many Barker codes have appeared in the gap during ranging and authentication and data regarding how many RUs have hit the gap, data determining the position of the Barker code(s) in the gap, and data ordering changes of position of the Barker code in the gap, data resulting from scanning the gap for additional unwanted pulses at the edges of the gap. This data is read by the CPU and used to compose messages for transmission by the transmitter on the control channel such as “no codes in gap-adjust your delays and try again”, “one code in gap”, “multiple codes in gap-enter contention resolution”, “move Barker codes x chips left or right”, “saw sequence xxxxxxx in gaps during authentication frames”, “no activity in gap during authentication interval-reexecute your contention resolution protocols” etc. 
     Equalization Training Process 
     Referring to FIGS. 54A,  54 B and  54 C, there is shown an embodiment of a process carried out by the RUs to carry out “training”. Training determines channel impairments and set coefficients into precode filters to predistort their transmissions such that their upstream data transmissions arrive at the CU undistorted. Training, in the preferred embodiment, also causes the modem to set the optimum transmitter power level and perform fine timing alignment. 
     Training is performed immediately after ranging and periodically thereafter. If the insertion loss, phase response and group delay were known for the channel and the effects of dispersion on the pulse shapes were known, intersymbol interference could be effectively controlled by the matched filters  761  in the CU receiver of FIG. 31 and 570 in the RU transmitter of FIG.  33 . However, even if these characteristics were known in advance, they tend to vary over time. Hence, in the preferred embodiment, an adaptive equalization process is performed to set variable coefficients in tapped delay line equalization filters to correct for the combined effects of residual distortion and noise caused by a dispersive and noisy channel. Prechannel equalization is performed in each RU and CU transmitter, and post channel equalization is performed in each RU and CU receiver in some embodiments. In the preferred embodiment, the training process is performed only for some filters in the system. Specifically, the CU precode equalization filter uses only averaged coefficients suitable for all RUs for transmissions in the downstream direction and the CE equalization filters in the RU receivers for downstream communications use only average coefficients found to be suitable for the average RU. Specific coefficients are computed for the SE circuits for each RU however after a training process similar to the process to be described below. This allows the equalized system to approach the ideal condition specified by the Nyquist criteria for distortionless transmission free of intersymbol interference so as to realize the full data carrying capacity of the channel. The adaptive equalization filters are tapped delay line filters in some embodiments with the tap delays equal to one chip time. In the preferred embodiment, the post channel filters are decision feedback equalizers. The equalization filters on both the transmit and receive side are embodied in precode equalization filter  563  in the transmitter of FIG.  32  and the FFE (feed forward) filter  764  and DFE (decision feedback) filter  820  along with least mean square calculation circuit  830  and difference calculating circuit  832  and FFE  765  in the receiver of FIG.  30 . 
     The equalization training process occurs in every RU as part of its startup sequence. The prechannel equalization process starts with establishment by the RU controller of default precoder coefficients, a default transmit power level (input on line  566  to the scaler amplifier  564  in FIG. 32) and a default fine timing alignment value in the preferred embodiment. Next, step  1101  in FIG. 53A is performed to transmit data on code #4. The RU uses only the first 8 CDMA codes during the equalization process. Step  1101  represents the process of transmitting any binary data bit sequence (preferably a pseudorandom sequence) to the CU using code #4 of the first 7 or 8 orthogonal spreading codes (the first 8 codes will be assumed for this example but it could be other numbers of sequential cyclic codes as well) to spread the data and using bipolar phase shift keying (BPSK). In step  1102 , the CU correlates the received data signal, after BPSK asynchronous demodulation, against each of the first 8 orthogonal, cyclic spreading codes. BPSK has only a two point constellation, so the CU is expecting to receive either of these two points from the correlation done between code #4 and the received signal if the ranging process has been done correctly. If the ranging process has not been properly fine tuned to put the RU&#39;s Barker code in the center of the gap, then the output data sent by the RU will be output from one of the other correlation processes which use one of the other 8 orthogonal, cyclic spreading codes. Each of the orthogonal, cyclic spreading codes is generated by shifting the code used during the previous chip time by one bit position. Therefore, each of the first 8 orthogonal, cyclic spreading codes is effectively different from its neighboring codes by one bit position and one chip time. If during the ranging fine tuning process, the Barker code was not exactly centered, the data transmitted by the RU will not be output by the correlation against code  4  but will be output by the correlation against one of the other codes depending upon how many chips away from the center of the gap the RU Barker code is found. Step  1104  is a test to determine if the data transmitted by the RU is output by the correlation against code #4. Step  1104  is preferably performed by checking the amount of code crosstalk by monitoring the demultiplexer memory. The CU also monitors the power level of the RU transmission by adapting the 4th tap of the FFE. If the training data did not come through purely on code #4 and crosstalk exists, it means the frame alignment is not perfect so step  1106  is performed to go back to the fine tuning process for ranging and center the RU Barker code in the gap. Step  1106  also symbolizes the process, in some embodiments, of computing a new power level and fine alignment value based upon measurements and sending them downstream to the RU in training. This process is repeated until the power level and frame alignment are within predetermined acceptable values of precision. The foregoing process of sending the proper power level to the RU from the CU may take the form of steps  1108 ,  1110  and  1112  in FIG.  53 AA. Step  1106  represents the process of telling the RU to go back to ranging and doing a fine alignment process in some embodiments, but in the preferred embodiment, it is not necessary to do the full fine tuning process detailed above for ranging since the CU knows exactly how far away from the center of the gap the data landed by virtue of which correlation computation put out the correct transmitted data. Therefore, if the code 3 correlation put out the transmitted training data, the transmit frame timing delay for this RU is off by one chip, and the CU sends a message to that RU telling it to move one chip toward the center. 
     In the preferred embodiment, the CU has an array of 8 correlators each of which correlates the received data using one of the first 8 orthogonal cyclic spreading codes. This arrangement is used for maximum speed. in other embodiments, a single correlator can be used on the buffered received data with the first 8 orthogonal cyclic codes being supplied during successive correlation intervals. In other alternative embodiments, the correlation can be done in serial or parallel in software. 
     Power Alignment 
     The equalization process also is used for power alignment. Power alignment of all the RUs is the process of setting their transmit powers so that their transmissions all arrive at the CU at approximately the same power level. This is important in preventing interference between the signals from different RUs as well as in allowing the CU receiver&#39;s detectors to properly interpret the QAM 16 constellation points which are distinguished from each other in part by their amplitude levels. This process is started with step  1108  in which the RU transmitter causes the gain of scaling amplifier  564  to be set to one. The CU receiver control circuitry then causes the initial gain level for code 4 to be retrieved from memory  796  and transmitted through control loop  781  to the gain control input  790  of G 2  amplifier  788 . This initial gain level set into G 2  amplifier  788  is an approximation of the proper gain level needed for this amplifier to allow slicer  800  to make proper decisions. Next, in step  1110 , the CU waits for its adaptive gain control circuitry to settle in at a gain level needed for low or no error interpretation of the BPSK modulated data being sent during the training interval. The adaptive gain control circuit is comprised of slicer  800  which outputs amplitude error numbers on bus  798  in FIG. 30 to control loop  781 . The control loop compares the amplitude error numbers to the current gain set on bus  790  and tries to adjust the gain number on bus  790  to minimize the slicer amplitude error. This process continues for a number of iterations by the end of which the gain of amplifier G 2  will have been set at a value which reduces the slicer amplitude error by as much as possible. Finally, in step  1112 , the CU takes this gain number on bus  790  (by reading the gain level on bus  790  from memory  796  via bus  797 ) and transmits it to the RU telling the RU to set that gain level as the gain of scaler amplifier  564  in FIG.  33 . The CU then sets the gain of G 2  amplifier  788  in FIG. 31 to one by writing a one into memory  796  as the gain level for code 4. 
     Since the overall gain of the system for code 4 is the gain of the RU transmitter amplifier times the gain of the CU receiver amplifier, the overall gain of the system does not change by swapping the gains. This power alignment process happens only for the RUs. Each RU, when it powers up, has its gain level aligned in this manner and will use that gain level for subsequent operation sending payload data until the power alignment is subsequently performed again. 
     Simultaneous Upstream and Downstream Equalization 
     Processing now moves on to the equalization process for both the upstream and downstream data path equalizers. The idea in downstream equalization is to set the tap coefficients of the FFE equalization filters in the RU receiver to values which equalize for channel impairments based upon errors observed in training data sent via the 8 training codes by the CU to the RU. The idea in upstream equalization is to set the tap coefficients of the precode filter in the RU transmitter to values which equalize channel impairments based upon information received from the CU receiver after training data is sent by the RU to the CU using the 8 training codes. The 8 training codes are the first 8 orthogonal, cyclic codes. They can and are used simultaneously in both the upstream and downstream directions (as are the rest of the codes) because the upstream transmissions are on a different frequency from the downstream transmissions. Although the flow chart of FIG. 54 (comprised of FIGS. 54A through 54C) shows the upstream equalization process occurring first, starting with step  1114 , both the upstream and downstream equalization processes are occurring simultaneously. FIGS. 54A through 54C show one alternative embodiment for an equalization and power alignment process. The preferred embodiment is discussed later herein. 
     Upstream 
     The first steps in the upstream equalization process are symbolized by steps  1114  and  1116  wherein, in step  1114 , the CU sends a message to the RU telling it to send some equalization training data (any data but preferably a pseudorandom PN sequence) to the CU using all 8 of the first 8 orthogonal, cyclic codes. In the embodiment shown in FIGS. 53A-53C, the CU requests that the RU send the training data using only one of the codes as a first transmission and then asks that the same data be sent using the other codes one code at a time or in small groups. With this method, if the RU&#39;s equalization filter coefficients are very far off the correct values, the transmissions by the RU in training will cause less intersymbol interference with payload data transmissions by other RUs that have already trained. However, in other alternative embodiments, the training data may be spread by all 8 of the first 8 codes and the results simultaneously transmitted. Step  1116  represents the process of sending the training data as spread by the first 8 codes to the CU either serially or all at once. When the RU sends training data, it sets tap coefficients of its precode equalization filter such as filter  563  in FIG. 33 to values that cause the precode equalization filter to not predistort the training data signal. 
     Step  1118  represents the process performed in the CU of receiving the equalization training data and using FFE equalizer  765 , DFE equalizer  820  and LMS circuit  830  to perform one iteration of tap weight (adaptive coefficient) adjustment for the tap weights of the FFE and DFE. Step  1120  represents the process of continuing to make tap weight adjustments on subsequent transmissions of equalization training data using the same first 8 orthogonal spreading codes until convergence is achieved when the error signals computed by difference calculation circuit  832  in FIG. 31 drop to near zero. 
     After convergence, step  1122  is performed wherein the CU sends the final tap weight coefficients from FFE  765  and DFE  820  to the RU. This can be done by the CPU  405  reading the tap weights from shared memory in which LMS stores them via bus  833  and transmitting them to the RU on the command and control channels. To calculate the new coefficients for the precode equalization filter  563  in the RU transmitter of FIG. 33, the old coefficients of the RU precode filter FFE and DFE equalization filter are convolved with the new coefficients FFE and DFE coefficients which the central unit modem symbol equalizer circuit converged on to derive new coefficients. These new coefficients are then set into the RU precode filter. 
     A block diagram of the structure of the precode equalization filter  563  in the RU transmitter of FIG.  33  and the FFE/DFE equalizer  764  in FIG. 31 is shown in FIG.  50 . The FFE/DFE circuits are comprised of a conventional FFE equalizer which receives the input data on bus  923  and outputs its results on bus  933  which is coupled to the plus input of a difference calculating circuit  925 . The minus input of the difference calculating circuit receives the output of a conventional DFE equalizer  929  via bus  931 . The output of the difference calculating circuit  925  on bus  927  is coupled to the input of the DFE equalizer  929 . 
     Finally, the CU, in step  1126 , sets the main tap of said FFE equalizer  921  in FIG. 50 to one and sets the side tap coefficients of the FFE equalization filter  921  and the DFE equalization filter  929  in FIG. 50 to zero equalizer  765  and  820 , respectively to one, for reception of upstream payload data. 
     In some embodiments, after the CU sends its tap weight coefficients to the RU in step  1122 , the process of steps  1114 ,  1116 ,  1118  are performed again for several iterations. The coefficients extracted from the CU adaptive equalizer on the second and subsequent iterations cannot be used directly in the precoder, since they were not produced from default precoder values. Instead, the new coefficients for use by the RU are computed as the convolution of the old precoder coefficients with the new adaptive equalizer coefficients. At each iteration, the CU evaluates the coefficients extracted from the adaptive equalizer, and when the errors have dropped below a predetermined threshold indicating that the RU is transmitting with adequate quality, then the training process is completed except for downstream training. 
     Downstream 
     Regardless of which upstream training embodiment is used, processing now proceeds to the downstream equalization training process. This downstream equalization process starts with step  1128  wherein the CU send equalization training data to the RU using all 8 training codes. Specifically, the CU sends a PN sequence simultaneously on 8 channels, each channel spread by one of the first 8 orthogonal, cyclic codes modulated using BPSK. Step  1130  symbolizes the process of the RU receiver receiving the equalization training data in multiple iterations and using the LMS circuit  830 , the FFE equalizer  765 , the DFE equalizer  820  and the difference calculating circuit  832  in FIG. 30 to converge on the proper FFE and DFE tap weight coefficients for the FFE equalizer  765  and the DFE equalizer  820 . After convergence, the RU CPU reads the final tap weight coefficients for the FFE equalizer  765  and the DFE equalizer  820  via bus  833  and calculates new tap weight coefficients for the FFE and DFE filters of the CE circuit  764  in the RU receiver of FIG. 30 by convolving the old CE filter tap weights with the FFE and DFE filter tap coefficients converged upon by the SE circuit during reception of multiple bursts of training data, and loads these newly calculated tap weight coefficients into the FFE and DFE filters of CE circuit  764  in the RU receiver of FIG. 30 via bus  844 , as symbolized by step  1132  of FIG.  53 C. The RU CPU  405  then sets the tap weight coefficients of the FFE  765  and DFE  820  to initialization values in this alternative embodiment so that they can reconverge as payload data is sent. In the preferred embodiment, the tap weights of FFe  765  and DFE  820  are maintained at their convergence values, and the tap weights of FFE/DFE  764  in the RU receiver of FIG. 30 are set to averages for all RUs. In some embodiments, step  1132  also involves sending a training status message indicating the success or failure of training, an indication of success being an implicit request to the CU to disable training transmissions on all channels. 
     The iterations of the processes symbolized by FIG. 53A through 54C occur every few milliseconds, and convergence occurs within a fixed amount of time. The processes are repeated every 2 minutes in some embodiments, and in other embodiments, retraining occurs both periodically and immediately upon activation of an RU&#39;s first and any subsequent timeslot assignments. In some embodiments, retraining occurs periodically at some lesser interval when an RU has no active timeslots and when a link quality monitoring process reports poor quality transmission. Retraining usually only takes 2 iterations for power and time alignment and one iteration for filter adaptation. In one embodiment, the CU does correlation between the coefficients currently being used by the RU and extracted coefficents on subsequent iterations. This means that the CU must know what coefficients the RU is currently using either by keeping a mapping of those coefficients in CU memory or by requesting them from the RU via the command and control channels. 
     After equalization training, the RU link to the CU is active, and the RU can begin receiving messages after receiving a hello message from the CU. The hello message gives the RU the CU&#39;s software revision number and the superframe offset number. The revision number allows the RU to check its software revision number for compatibility, and the superframe offset number is set into a SFDOR register in the RU receiver time base for use in correctly reproducing an external time division multiplex stream superframe signal at the appropriate spot in the data stream so that external devices that depend upon the superframe signal can correctly interpret the TDM data. 
     Carrierless Modulation 
     Referring to FIG. 42, there is shown the preferred form of carrierless modulators used in the RU and CU transmitters. In the modulator of FIG. 23, multipliers  429  and  435  are used to multiply the incoming data times the local carrier sine and cosine signals. The result is two orthogonal RF signals bearing the inphase and quadrature information. 
     This same result can be achieved in a substantially different way by using Hilbert transform filters and carrierless amplitude and phase modulation. In the preferred form of modulator  507  shown in FIG. 42, the multipliers  429  and  435  and local oscillator  425  and phase shift circuit  439  in FIG. 23 are eliminated thereby resulting in a less expensive, less complex modulator that achieves the same result as the modulator of FIG.  23 . Specifically, shaping filter/modulator  507  of FIG. 42 receives inphase (real) and quadrature (imaginary) digital inputs (or analog) on buses  568   r  and  568   i . Although, buses  568   r  and  568   i  are shown in FIG. 42 as originating at the results array for clarity of illustration, in the preferred RU and CU transmitters of FIGS. 32 and 33, they actually originate from the output of the scaling circuit  564 . In some embodiments, the scaling circuit  564  and the precode equalization filter  563  can be eliminated where higher error rates or less payload capacity can be tolerated. 
     The Fourier spectrum of the baseband, orthogonally code division multiplexed data on bus  568   r  is shown as a constant amplitude spectrum  1138  of amplitude A r  on the real axis in FIG.  43 . The Fourier spectrum of the baseband, orthogonally code division multiplexed data on bus  568   i  is shown as a constant amplitude spectrum  1140  of amplitude A i  on the imaginary axis in FIG. 43 The direct sequence spread spectrum techniques employed in the transmitters according to the teachings of the invention has the effect of spreading the energy of the signals represented by the information vectors in frequency from minus infinity to plus infinity at a constant amplitude. Because any 6 mHz wide section of the spectrum of FIG. 43 can be selected with a passband filter and all the channel data therein recovered, this fact is employed to simultaneously carry out carrierless amplitude and phase modulation as well as filtering to satisfy the Nyquist criteria in shaping filter/modulator  507 . To do this, two shaping filters H R    1134  and H I    1136  in modulator  507  are coupled to receive the signals on buses  568   r  and  568   i , respectively. Filter  1134  has its filter characteristics set (programmable by CPU  405  in some embodiments) to establish a “squared-raised cosine” passband filter characteristic  1142  in the real plane of the frequency domain shown in FIG.  44 . The passband filter characteristic has a bandwidth of 6 mHz and is centered on an intermediate frequency Fc which is established at a frequency which can be easily and conveniently achieved in a digital filter. The output signals of the filter are ultimately sent to digital-to-analog converter  576  in FIGS. 32 and 33 and from there to frequency translator up/down converter  577 . The function of the up/down converter  577  is to raise the frequency to a frequency in the band devoted to digital data communication and assigned to upstream or downstream communications as appropriate to implement the CATV or cellular system supplemental services on the shared transmission media  412 . 
     Filter  1136  also has a “squared-raised cosine” passband filter characteristic  1144 , but its filter characteristic is located in the imaginary plane of the frequency domain shown in FIG.  44 . The passband filter characteristic has a bandwidth of 6 mHz and is centered on an intermediate frequency Fc which is easy to attain in digital filter design. To insure orthogonality between the real and imaginary data output signals on buses  1146  and  1148 , the transfer function of filter  1136  is the Hilbert transform of the transfer function of filter  1134 . 
     When the baseband spectra of FIG. 43 for the real and imaginary signal components are passed through filters  1134  and  1136 , the resulting Fourier spectra of the digital data on buses  1146  and  1148  are as shown in FIG.  44 . These spectra contain all the encoded information from the real and imaginary information vectors encoded by the orthogonal code multiplexer  527 . These digital signals on buses  1146  and  1148  are summed in summing circuit  1150 . 
     Referring to FIG. 54, there is shown a block diagram of an alternative embodiment of a system employing simple CU and RU modems according to the genus of the invention and using forms of modulation and multiplexing options for downstream data including SCDMA, DMT, TDMA, FDMA, etc. The system comprises a CU modem  1160  coupled by an HFC (hybrid fiber coax) or wireless transmission media such as a cellular or satellite radio transmission system  1162  to one or more RU modems  1164 . The purpose of the CU modem is to provide a multiple-user and/or multiple-source simultaneous digital data communication facility over a limited bandwidth channel such as 6 megahertz to one or more remote unit modems coupled to the CU modem by a shared RF transmission media. 
     The CU modem transmits data in the downstream direction toward the RU modems using a transmitter  1170  that uses digital data to modulate one or more radio frequency carriers that are transmitted over the media  1162  after frequency conversion by up/down frequency converter  1174  to the proper assigned downstream channel frequency. The transmitter can use any modulation and any multiplexing scheme which can effectively transmit a master clock reference and a master carrier reference signal as well as payload data to the RU modems. The clock and carrier references may be transmitted either in-band or out-of-band. Data is transmitted in frames which the RU receiver detects. The RU transmitter achieves frame synchronization by the ranging processes described elsewhere herein or by any other means. Examples of multiplexing schemes that will work for the downstream direction CU transmitter are TDMA, synchronous TDMA, FDMA, Inverse Fourier, SCDMA or DMT (digital multitone transmitter). Any compatible modulation scheme can be used. Any of the conventional transmitters described in the treatises incorporated by reference herein will suffice for the CU transmitter, but an SCDMA transmitter is preferred. Non-SCDMA multiplexing schemes can be used in the downstream direction because the noise and interference problems are less severe than in the upstream direction. 
     The definition of “in-band” transmission of the clock and carrier is that one or more channels which would otherwise be used to transmit payload data are dedicated to transmitting the clock and carrier signals. The definition of “out-of-band transmission is that a separate carrier or some other subchannel/sideband etc. modulation scheme is used to transmit the clock and carrier information so that no timeslot or packets that could be used to send payload data is used to send clock and carrier information. In the embodiment of FIG. 54, the master clock signal is generated by master clock  1176  and the master carrier reference signal is generated by oscillator  1178 . This master carrier signal is modulated by transmitter  1170 . 
     The CU transmitter has a framing/addressing/packetization circuit  1166  which functions to receive payload data at an input  1168  and organizes said data into frames and addresses the data to the proper destination RU modem and the proper peripheral device coupled to that RU. The manner in which this is done is not critical to the invention so long as the downstream data is organized into frames since the upstream data is transmitted by SCDMA. The CU transmitter&#39;s framing addressing circuit  1166  can have the structure and operation of the framing circuit  400  in FIG. 8 if the transmitter  1170  is an SCDMA or DMT transmitter. If the transmitter  1170  is, for example, a TDMA or synchronous TDMA transmitter, the framing/addressing circuit  1166  organizes the data into frames and places data bound for specific RU modems into timeslots assigned to those RUs. The data in these timeslots of each frame assigned to a particular RU will includes header bits which tell the RU modem to which particular peripheral or other destination the data in these timeslots is addressed and may include other information such as packet delimiters which define the start and stop of each packet destined to a particular RU or peripheral or may include byte counts etc. which tell the RU how many timeslots of data to collect for a complete packet destined for a particular destination coupled to that RU. Basically, the function of the framing/addressing/packetizing circuit  1166  includes organizing the payload data such that information as to which RU and peripheral each payload data byte is directed to can be determined. 
     The CU receives upstream radio frequency signals using an SCDMA receiver  1172 . The function of the SCDMA receiver is to extract the payload data from the upstream RF signals. In the preferred embodiment, the SCDMA CU receiver receives the master clock and master carrier signals on lines  1181  and  1180  and periodically recovers the clock and carrier from the RU signals using the preamble data and Barker codes transmitted by the RU in the manner previously described. This upstream payload data is multiplexed by an SCDMA transmitter in the RU modem using orthogonal, pseudorandom spreading codes. Then a suitable modulation scheme such as QAM is used to control one or more characteristics of one or more RF carriers to generate the upstream RF signal. The CU receiver  1172  can have the structure of the receivers of FIG. 28 or  32  or the more basic, but lower performance structure of FIG. 54 or any other code division multiplexed receiver structure which is compatible with the ranging processes defined herein. In the preferred embodiment, the RU transmitter uses clock and carrier signals which are synchronized to or at least phase coherent with the master clock and master carrier signals used by the CU transmitter and the RU receiver. In these embodiments, the CU receiver  1172  does not include tracking loops to continuously track the phase and frequency of the clock and carrier signals used by the RU transmitter to generate the upstream signals. In these embodiments, each RU transmitter send preamble data prior to sending upstream payload data. This preamble data from each RU is used by the CU SCDMA receiver  1172  to determine the phase differences between the clock and carrier signals used by each RU transmitter and the master clock and master carrier signals used by the CU transmitter. These phase differences are detected once for each RU and stored in a memory location dedicated to that RU. These phase differences are updated each time the RU transmits preamble data. The SCDMA receiver in the CU is informed by CPU  1194  each time preamble data is being received by activation of a PREAMBLE DATA signal. This signal is sent to the SCDMA receiver  1172  by a line  1195 . FIG. 54 is also intended to symbolize embodiments wherein the RU transmitter uses its own clock and carrier signals which are unrelated to the CU clock and carrier signals and wherein the SCDMA receiver includes tracking loops to continuously track the phase and frequency of the clock and carrier signals used by each RU. In these latter embodiments each RU sends preamble data prior to sending upstream payload data. This preamble data is sufficient for the tracking loops in said SCDMA receiver  1172  to lock onto the phase and frequency of the clock and carrier signals used by that RU for the time when upstream payload data is being received from that RU. 
     The RU modem  1164  has the following structure. A receiver  1190 , having a demodulator and detector compatible with the type of modulation performed in the CU transmitter, is coupled to the transmission media  1162 . The function of the RU receiver is to receive downstream RF signals transmitted by the CU transmitter, recover the master clock and master carrier of the CU, and synchronously extract the downstream payload data transmitted by the CU. The RU receiver also extracts any management and control data transmitted by the CU to coordinate the process of achieving frame synchronization and coordinates with the CPU  1204  and the RU SCDMA transmitter  1210  to carry out the ranging process or whatever other process is used to achieve frame synchronization. The RU receiver recovers the master clock and the carrier used by the CU transmitter in the manner described above or in any other way known in the prior art. The recovered master clock signal is distributed on bus  1214  to all RU circuits that need it including the SCDMA transmitter  1210  in embodiments that eliminate the tracking loops in the CU receiver. The recovered carrier signal is distributed by receiver  1190  on bus  1216  to all circuits that need it including the SCDMA transmitter  1210  in embodiments that eliminate the tracking loops in the CU receiver. The preferred method of recovering the clock in the RU is by encoding the clock into Barker codes sent during every gap by the CU and using a gap monitor/frame detector circuit like that shown in FIG. 34 to generate clock steering signals to keep an RU local clock oscillator in synchronization with the master clock  1176 . The preferred method of recovering the carrier is by dedicating one channel or timeslot to a pilot tone that defines the phase and frequency of the master carrier generated by the master carrier local oscillator  1178  in the CU and monitoring that channel to generate steering signals to keep a local oscillator in the RU synchronized. 
     The RU receiver  1190  can have the structure of the receivers described in FIGS. 28 or  31  and  34  as well as alternatives and functional equivalents thereof mentioned herein or known to those skilled in the art, or it can have the structure of conventional receivers described in the treatises incorporated by reference herein. The only requirement is that whatever structure the RU receiver has, it must be capable of decoding and extracting the downstream payload and management and control data transmitted by the CU transmitter. The extracted payload data is output on bus  1216  for use by peripherals and interfaces to other networks or processes represented by block  1218 . 
     An RU transmitter  1210  receives payload data on bus  1220  from the peripheral devices or processes and organizes that data into frames of the same size as the CU frames. The data so framed then has its Fourier spectrum spread by the transmitter over a bandwidth much larger than said data originally had by orthogonal code division multiple access encoding or by performing an inverse Fourier transform operation. In some embodiments RU transmitter  1210  can be a synchronous TDMA transmitter. If code division multiple access is used, the spread spectrum data is then modulated onto one or more radio frequency carrier signals using a suitable modulation scheme such as QAM 16  as described elsewhere herein. The process of organizing the upstream data into frames, spreading the spectrum of each frame of data and using the spread spectrum data to modulate one or more RF carriers is done synchronously with the CU using the master clock and master carrier signals recovered by receiver  1190  and output on buses  1214  and  1216 , respectively. In some embodiments the RU local carrier reference signal on line  1216  is a different frequency but phase coherent with the master carrier. The resulting RF signals are output on line  1224  to an up/down frequency converter  1226  where the frequency is converted to the designated frequency of an upstream frequency band, usually 6 mHz in width, and then the resulting signals are output on line  1228  to the transmission media  1162 . Therefore, frequency division multiplexing for the upstream and downstream traffic is employed to share the transmission media between upstream and downstream channels. Those skilled in the art will appreciate that the system of the invention uses a combination of time division multiplexing, frequency division multiplexing and code division multiplexing to achieve high-performance, multiple-user, multiple-source bidirectional digital data traffic in a distributed communication system. 
     Frame synchronization is achieved, in the preferred embodiment, by the trial and error process of adjusting the transmit frame timing delay described elsewhere herein and then transmitting a unique code such as a Barker code which the CU receiver can detect using that transmit frame timing delay following detection of the CU gap by frame detector in RU receiver  1190 . The CU modem includes a gap monitor circuit  1192  (shown separately but which could be inside SCDMA receiver  1172 ) that functions to monitor the CU guardband or other interval included in each frame to which the RU transmitters are trying to synchronize to determine if one or more RU Barker codes have been received. The gap monitor circuit can have the structure shown in FIG. 34 or any other structure that can: determine when the unique code of an RU has been received; determine if more than one code from an RU has been received in the gap; detect how far away from the center of the gap the received Barker code is; and, provide status information on bus  1196  to a computer  1194 . The status information tells the CPU  1194  whether a Barker code has been received, if more than one Barker code has been received, and, if only one Barker code has been received, and how far away from the center of the gap the received Barker code is. Although a computer is preferred for circuit  1194 , other circuits to perform this function such as gate arrays, state machines etc. may be used to generate the management and control data on bus  1198  which informs the RUs of information they need to achieve frame synchronization. Hereafter, circuit  1194  will be referred to as a computer. The same is true of computer  1204  in the RU. The computer  1194  then generates management and control message data on bus  1198  which are presented at one input of a switch  1200  the switching state of which is controlled by computer  1194  (or other timing logic) to select the data on bus  1198  during the interval for transmitting data from timeslots devoted to management and control messages. Those skilled in the art will appreciate that a switching multiplexer like MUX  1200  need not be used and any other known data transfer circuit or process to get data from one process to another such as shared memory etc. may be used to get the management and control data transmitted at the proper time. For example, the management and control data may be stored in specific locations of a shared address space of a memory which also stores the output data from the framing circuit  1166 , and the transmitter may have a computer or state machine which accesses the memory at the proper times to send the data assigned to various timeslots including the management and control data. 
     The RU receiver  1190  receives these management and control messages and passes them on bus  1202  to a computer  1204  which uses the management and control data to control the ranging process carried out by said SCDMA transmitter  1210  and Ranging Generator  1206  and for other purposes. The gap monitor circuit in receiver  1129  supports the CU gap acquisition process by locating the time of each CU frame gap. This gap monitor circuit listens for Barker code data transmitted by the CU during every gap, usually by correlating received energy against the known Barker code data pattern, and sends gap acquisition data detailing the receipt of correlation pulses and the relative times of their occurrence to computer  1204  via bus  1202 . In some embodiments, the gap monitor circuit is structured like the frame detector of 34 and uses the Barker code transmitted by the CU to recover the CU master clock by an early-late gating technique. 
     Computer  1204  or other control circuitry uses this gap acquisition data to determine the time of receipt of the Barker code thereby establishing a frame boundary reference for the receiver to aid it in demodulating, decoding and deframing the received data and a reference time from which to start the transmit frame timing delay. Specifically, the computer  1204  uses the receive frame timing reference during the ranging process to establish the starting time against which to measure a trial and error value for the transmit frame timing delay value T d , and then sends this transmit frame timing delay value T d  on bus  1212  to the RU transmitter  1210  to control the delay between the time when a frame of downstream data arrives from the CU transmitter, and the time the RU transmitter  1210  sends the same frame back to the CU receiver with new upstream data therein. During the ranging process, the value of T d  is varied experimentally during successive Barker code transmissions until management and control data is received by the RU indicating that the Barker code has been centered in the CU frame gap thereby achieving frame synchronization. Once frame synchronization has been achieved, the computer  1204  freezes the value for T d  thereby causing the SCDMA transmitter to send its frames in frame synchronization with the CU frames and frames transmitted by all other RUs. 
     The frames all have numbers and can be visualized in the following manner. A downstream frame travelling from the CU transmitter to the RU receiver is like a bus with a load of particular people, the people being the payload and management and control data in the frame. When that bus arrives at the RU, the people are unloaded, and a new set of people get on representing the payload and management and control data the RU wants to sent to the CU. After delay T d , the bus leaves the RU and travels back to the CU. The different channels of data can be visualized as different buses each destined for a different RU although in reality, they are data from different timeslots in the same frame which is received by all RUs. Frame synchronization is the process of setting the value of T d  properly in each RU after a trial and error process so that the buses from each RU travelling toward the CU all arrive at the same time. 
     As a further operation in achieving frame synchronization, the computer  1204  also enables a ranging generator circuit  1206  via signals on a bus  1208  and passes messages to the ranging generator to control its operation. The ranging generator  1206  functions to generate and send to the RU transmitter  1210  data defining the Barker code for transmission during a ranging process, the power level for transmission and the unique on-off morse code authentication signature sequence that is used to identify each particular RU during the ranging process. 
     Computer  1204  also generates and sends management and control data to the RU SCDMA transmitter  1210  via bus  1212 . This management and control data can include requests to start ranging, requests for more bandwidth, messages relinquishing bandwidth etc for various species within the broad genus of the invention. 
     Referring to FIG. 55, there is shown a block diagram of a simple form for the SCDMA receiver in the CU of block  1172 . This embodiment has a demodulator  1230  which receives the modulated RF signals on line  1232  and the master carrier reference signal on line  1180  from the master carrier local oscillator  1178 . The demodulator also may receive the master clock signal on line  1234  in some embodiments where the output signal is converted to digital samples and output as a baseband signal on bus  1236 . These baseband signals are coupled via bus  1236  to the SCDMA demultiplexer  1238 . The demultiplexer  1238  multiplies the results vectors times the transpose of the code matrix that the information vectors were multiplied by in the SCDMA multiplexer of the RU transmitter. This despreads the Fourier component power spectrum and results in signals output on bus  1240  in frame format in accordance with framing information received on bus  1244  from the CU transmitter. This framing information defines the CU frame times, but the RU data frames are arriving synchronously with this frame timing. 
     The signals on bus  1240  are corrupted by noise and impairments that degrade the upstream channel. In addition, the phase and amplitude errors, for each RU need to be removed. Accordingly, the detector  1246  includes a G 2 /rotational amplifier and a memory for storing gain and phase adjustment numbers, as well as a slicer. When preamble data is being received, the CPU so informs the detector by signals on bus  1241 . This bus is also used to inform the detector which timeslot is being received so that the detection can retrieve the proper gain and phase connection factors. The corrupted phase and gain adjusted information signals can then be processed by a conventional decoder in detector  1246  to determine the actual constellation points that were sent. In the preferred embodiment, the decoder is a Viterbi decoder and FFE and DFE equalization is optionally used along with Trellis modulation at the RU transmitter to improve throughput, decrease error rate and improve signal to noise performance. 
     The information vectors resulting from the decoding process are then output on bus  1248  to a deframer circuit  1250  which functions to reassemble the original payload data streams, ATM cells, LAN packets or TCP/IP packets and management and control messages from the information vectors in each frame. The payload data is output on bus  1252  to the peripherals and interfaces to the phone network, internet etc. The management and control data are output on bus  1254  to the CPU  1194  for use in processing such as assigning channels and dynamic bandwidth management in species that implement these functions as opposed to fixed channel assignments. 
     An RU SCDMA receiver could be structured like the receiver of FIG. 55 in embodiments using SCDMA downstream transmission. However, instead of using the master clock and master carrier reference signals of the CU, recovered clock and recovered carrier signals would be used in the RU receiver. These recovered clock and carrier signals could be generated by the same circuitry in FIG. 8 that perform these functions. 
     Referring to FIG. 56, there is shown a block diagram of a simple RU spread spectrum transmitter which could be used to implement block  1210  in FIG. 54. A framer circuit  1260  receives payload data on bus  1220  from the peripherals and organizes it into frames of the same size as the CU frames. The framer can have the structure of framer in FIG. 9 or some other structure that organizes the frames of information vectors differently. The framer receives frame timing information in the form of frame sync and super frame signals on bus  1262  from time base  1265 . Time base  1265  has a tracking loop therein and receives a clock synchronization steering signal on bus  1264  from the RU receiver gap detector, and uses this signal to keep its local clock in VCXO synchronization with the master clock in the CU. A synchronized chip clock reference signal is distributed on bus  1266  to all circuits in the transmitter that need it. The time base  1265  also receives a receive frame timing reference on bus  1268  from the computer/control circuit  405 . The control circuit  405  receives gap acquisition data on bus  1270  from the gap monitor circuit in the RU receiver and uses this gap acquisition data to determine when the CU frames arrive. This information is also used to generate the transmit frame timing delay T d  on line  499 . 
     The framer circuit  1260  outputs information vectors on bus  1272  to one input of a switch  1274 . The other input bus  1276  of this switch is coupled to the computer  405  and carries management and control data generated by the computer. The switch  1274  has a single output bus  1276  which is coupled to the data bus input of a buffer memory  1278 . The buffer memory serves to store the final information vectors which will be input on bus  1280  to the spectrum spreading multiplexer  1282 . The multiplexer  1282  functions to spread the Fourier spectrum of the data on bus  1280  over a much larger bandwidth than the data originally had. In the preferred embodiment, the multiplexer  1282  is a direct sequence code division multiplexer like those described elsewhere herein which carries out matrix multiplication between the information vectors and a plurality of pseudorandom, orthogonal codes, which are preferably cyclic codes. At least one code is assigned to each RU that has payload data to send, and in the preferred embodiment, multiple codes may be assigned when more bandwidth is needed by an RU. The number of codes assigned to an RU can be dynamically varied by exchanges of messages between the RU and CU via management and control channels. The computer  405  receives code assignment messages from the CU as well as other management and control data which supports, for example, the ranging process, via bus  1288  from the RU receiver. 
     The computer  405  carries out the assignment of codes per instruction from the CU (or frequency components in FFT and DMT embodiments) by controlling read pointer addresses on bus  1284  coupled to framer circuit  1260  and write pointers on bus  1286  coupled to buffer  1278 . The read pointers control the addresses in the framer circuit from which data is read for output on bus  1272 . The write pointers control the addresses to which data on bus  1276  is written into buffer  1278 . Since the contents of any particular address in buffer  1278  always get multiplied by the same code, by controlling these write pointers, the computer can implement the code assignments and put management and control data in the correct channels if specific channels are assigned for management and control data. The embodiment of FIG. 56 includes the capability to shuffle codes pseudorandomly (or frequency components in DMT embodiments) by pseudorandomly altering the write pointers to place data from specific channels into pseudorandomly assigned locations that will get multiplied by different codes. 
     The transmitter of FIG. 56 can also implement inverse FFT and DMT multiplexing by using an appropriately structured spectrum spreading circuit  1282 . To implement inverse FFT embodiments, block  1282  represents a process to calculate the inverse Fast Fourier Transform using as the different frequency component magnitudes magnitudes represented by individual information vector elements on bus  1280 . The inverse FFT embodiment uses the same information vector elements to define the magnitude of the same frequency components during each frame. A DMT or digital multitone system is like an inverse FFT system but alters (either pseudorandomly or sequentially) the frequency components assigned to each vector element from time to time. To implement a DMT embodiment, block  1282  performs the inverse Fast Fourier Transform, and computer  405  controls the read and write pointers to alter the frequency components assigned to each information vector element. 
     Whatever the spectrum spreading process carried out by block  1282 , the resulting data is output on bus  1290  to one input of a switch  1292 . The other input bus to this switch is coupled to receive Barker code data on bus  1294  from a Barker code generator/authentication sequence generator  1296 . The switch functions to selectively couple the data on bus  1294  to the input of a modulator  1298  via bus  1300  when the CPU changes the state of the switch via control line  1302  at time delay T d  after the CU gap is detected during ranging. Generally, the switch is controlled to send the Barker code data on bus  1294  to the modulator  1298  during the ranging process and to send payload data on bus  1290  to the modulator during normal operation after frame synchronization has been achieved. Computer  405  controls the Barker code generator  1296  via data on bus  1304 . 
     The payload data extraction process is done synchronously in the CU and RU modem receivers. “Synchronously” as that word is used in the claims means the following forms of synchronization are practiced in the RU receiver, CU SCDMA receiver and the RU SCDMA transmitter. The RU transmitter uses the recovered master clock and master carrier reference signals recovered by the RU receiver to drive its digital circuitry and modulator in synchronism with the CU master clock and master carrier. Coherent detection is performed in the CU SCDMA receiver using the master carrier signal on line  1180  and a rotational amplifier or using a recovered carrier from either an in-band source like the pilot channel described elsewhere herein or some out-of-band source and driving the demodulator with the recovered carrier. In the preferred embodiment, the CU&#39;s SCDMA receiver uses its own master clock and master carrier without recovering either from the signals transmitted by the RU. These signals plus information derived from each RU&#39;s preamble data provides knowledge in the CU SCDMA receiver of the RU&#39;s SCDMA transmitter carrier phase and frequency. An RU SCDMA or other type of receiver recovers the master carrier reference from, for example, the pilot channel transmitted by the CU and recovers the master clock reference from the Barker codes sent by the CU during the gaps of every CU frame. Those recovered master clock and master carrier signals are used for synchronous demodulation in the CU and to synchronize the detector in the RU receiver and are also used by the RU SCDMA transmitter. Frame synchronization is also part of the synchronization implied by the term “synchronously” in the claims. 
     Referring to FIG. 57 there is shown a block diagram of a synchronous TDMA system for bidirectionally communicating digital data over any transmission media including hybrid fiber coax using FDMA upstream and downstream channel separation so as to not interfere with other services such as cable television programming sharing the HFC. The CU modem  1380  receives a TDMA stream of data from higher level software layers, peripherals or other interfaces such as a T1/E1 line, and synchronizes its own master clock  1384  from signals on the TDMA bus  1382  that define the frames of timeslots thereon. The TDMA stream on bus  1382  is received by a CU TDMA transmitter  1386  which also receives a master clock signal on bus  1388  and a master carrier reference signal on bus  1390  from a master carrier reference oscillator  1392 . The TDMA transmitter receives the frames of data and modulates the data from each timeslot of each frame onto one or more carrier signals supplied by the master carrier oscillator  1392  using any modulation scheme which can transmit the master clock and a carrier reference signal to the RU modem either in-band or out-of-band. Examples of such modulation schemes include QAM, QPSK etc. For example, one or more time slots may be devoted to sending data encoding the master clock signal and master carrier reference. Alternatively, one timeslot can be devoted to carrying the master carrier as a pilot channel signal, and the master clock may be sent embedded in Barker codes sent in gaps between frames. In alternative embodiments, the downstream data can be transmitted by a CU transmitter  1386  which uses any other multiplexing scheme other than TDMA. 
     The modulated RF signals are output on line  1394  to an up/down frequency converter  1396  which converts the frequency thereof to a downstream frequency which will not interfere with other services sharing the transmission media  1398  such as cable TV programming fed into the media from bus  1400 . The frequency converted signals are output on line  1402 . Frequency conversion is optional if the master carrier in the CU modem can generate a carrier at the desired downstream frequency and the upstream channel can be some frequency which can be synchronized to the downstream frequency such as a harmonic. An RU modem  1404  receives the downstream data on line  1408 . A TDMA receiver  1406  coupled to line  1408  recovers the master clock and master carrier reference signals using any conventional circuitry or the circuitry and methods disclosed earlier herein in the STDMA embodiments. The TDMA receiver  1406  outputs the recovered clock signal on line  1410  and outputs the recovered carrier signal on line  1412 . The recovered payload data is reassembled into a TDMA data stream and output on bus  1414  to peripherals or other interface processes. 
     Those peripherals or other interface processes also supply a TDMA input data stream on bus  1416  to an RU synchronous TDMA transmitter  1418 . This transmitter receives the recovered clock and recovered carrier signals on lines  1410  and  1412 , respectively, and synchronously organizes the TDMA input data on bus  1416  into timeslots in TDMA frames having the same duration as the CU frames. These frames are then modulated onto one or more carrier signals using the same (or different) modulation scheme used by the CU transmitter, and the frames of modulated RF signals are transmitted to the CU in frame synchronization with the CU. That is, the RU frames are transmitted from the RU transmitter with a transmit frame timing delay set for this particular RU&#39;s position in the system relative to the CU such that the frames transmitted by the RU arrived at the CU aligned with the CU frame boundaries. All RU modems in the system have their transmit frame timing delays set for their particular positions on the network so that all their frames arrive at the CU aligned with the CU frame boundaries. All RU&#39;s also send preamble data prior to sending payload data for use by the CU in determining the phase error for that particular RU. This preamble data is used in the manner described above for the SCDMA embodiments to find the phase error. 
     The modulated RF data output by RU TDMA transmitter  1418  is coupled on line  1420  to an up/down frequency converter  1422  that functions to change the frequency of the upstream channel to a frequency that is far enough removed from the downstream channel frequency and from the cable TV programming so as to not interfere therewith. 
     The upstream data is then transmitted via line  1424  and the transmission media to a CU TDMA receiver  1426 . This receiver receives a master clock signal on line  1428  from the master clock oscillator  1384  and receives the master carrier signal on line  1430  from the CU&#39;s master carrier reference oscillator. The CU TDMA receiver  1426  also receives a PREAMBLE signal on line  1441  from the CU CPU  1442 . This signal is activated when preamble data is being received. The CPU also supplies receiver  1426  with an RU ID signal on line  1439 . This signal tells the receiver  1426  which RU&#39;s data is being received in the timeslot currently being received. The CPU keeps track of which RU&#39;s data is being received with the help of data received on line  1440  from receiver  1426 . The receiver  1426  includes a frame detector which detects the Barker codes transmitted during the ranging process by the RU&#39;s. The TDMA receiver&#39;s frame detector can have the same structure as the frame detector previously described herein. The data on line  1440  tells the CPU when each RU has achieved frame synchronization. From that point, the CPU knows that the RU frames are coincident with the CU frame boundaries. Line  1440  also carries data received from the RU&#39;s requesting bandwidth. In response to these requests, the CPU would assign one or more timeslots to the requesting RU. These assignments would be conveyed to the RU&#39;s by management and control messages generated by the CPU  1442  and sent to the CU transmitter via bus  1444 . The CPU keeps track of which timeslot is being received with the help of master clock data on line  1443 . The CPU then looks up the RU assigned to each timeslot and sends that information to the TDMA receiver  1426  as the RU ID signal on line  1439 . 
     The CU TDMA receiver  1426  recovers the payload data from the modulated upstream signals and reassembles the payload data into a TDMA output data stream on bus  1432 . 
     The TDMA transmitters and receivers in this system can be conventional, but the RU TDMA transmitter must be able to delay transmission of its frames by a variable transmit frame timing delay so that its frames arrive in frame synchronization with the frame boundaries of the CU. Any ranging process described herein or any other known ranging process can be used to achieve this frame synchronization. If any of the trial and error class of ranging processes described herein is used, computer  1434  in the RU modem sets an initial transmit frame timing delay either at its own initiative or upon receipt of a ranging solicitation message from the CU via a management and control data path  1436  from the receiver  1406 . This initial delay value is sent to the RU transmitter via bus  1438 . The CU receiver assists in the ranging process by sending data regarding what signals from the RUs it found in the frame gaps if gaps are used or what RU ranging signals were detected over the frame interval via bus  1440  to CU computer  1442 . The CU computer  1442  sends feedback ranging data to the RU via bus  1444  coupled to the CU transmitter  1386 . 
     In the class of embodiments where the CU does the ranging process for the RU by determining how much the RU must move its ranging pulse to achieve frame synchronization and so instructing the RU, bus  1440  still carries data regarding what ranging pulses the CU receiver saw. However, computer  1442  then figures out how much delay the RU needs to add to or subtract from its transmit frame timing delay by calculating the total turnaround time and then sends a message via bus  1444  to the RU so instructing it. This message reaches RU computer  1434  via bus  1436 , and the computer  1434  sets the instructed delay via bus  1438 . Any other ranging process that can achieve frame synchronization other than the ones described herein will also suffice to practice this particular embodiment. 
     Active Bandwidth Management 
     All of the transmitter embodiments disclosed herein can utilize an active bandwidth management process carried out by bidirectional message traffic between the remote units and central unit over the management and control channels. Remote units can request more or less bandwidth on a first-come, first-served basis or the RU&#39;s can request reserved bandwidth, i.e., bandwidth that has been reserved to each RU but which can be loaned to other RU&#39;s until the RU for which the bandwidth is reserved need it. The central unit can evaluate RU privileges for bandwidth reservation, privileges, etc., and arbitrate conflicting requests for reserved bandwidth or more bandwidth and then award bandwidth in accordance with the results. The CU then sends downstream management and control messages telling each remote unit which codes have been assigned to carry its traffic during specified frames. 
     Upstream Time Alignment Algorithm 
     Section 1.1 Time Alignment Procedure 
     Time alignment is the procedure by which fine delay adjustments are made to provide exact frame synchronization of all RU&#39;s to exactly center the RU Barker codes in the gaps. (Coarse frame synchronization is accomplished by ranging). Every RU undergoes an initial upstream training process to achieve coarse frame synchronization, exact centering of its Barker code in time alignment, power alignment to solve the near-far problem, and equalization to predistort transmissions to minimize the effects of channel impairments. Time alignment fits into the overall upstream training procedure as follows: 
     For initial training: 
     1. Ranging by any of the processes previously described 
     2. Time alignment 
     3. Power alignment 
     4. Time alignment 
     5. Power alignment 
     6. Equalization 
     7. Repeat steps 4-6 N times 
     For periodic training: 
     1. Time alignment 
     2. Power alignment 
     3. Equalization 
     Section 1.2 presents the requirements for one embodiment of time alignment, with input requirements dictated by ranging and output requirements dictated by system performance. Section 1.3 gives a detailed description of the time alignment algorithm of one embodiment with Section 1.4 giving alternative enhanced embodiments to the algorithm of Section 1.3. 
     1.2 Time Alignment Requirements 
     The ranging procedure is capable of aligning the upstream Barker code transmission to within +/−1 chips of the center of the gap and aligning the power to within a number of dB which is acceptable for the system performance criteria. Time alignment must operate within these constraints for this embodiment. 
     The demultiplexer and equalizer circuits require fine time alignment to within +/−1 high-speed clock in this embodiment. 
     Initial time alignment must be accomplished within a number of seconds in this embodiment which is acceptable for the system performance criteria. 
     Periodic time alignment must be accomplished within a number of seconds in this embodiment which is acceptable for the system performance criteria. 
     Time alignment phase shifts must be compensated in this embodiment. 
     1.3 Algorithm Description 
     The time alignment algorithm has two main components: coarse time alignment and fine time alignment. Coarse alignment can begin with offsets of up to +/−8 chips and align to within +/−½ chip. Fine alignment begins with offsets of up to +/−1 chip and aligns to within +/−1 high-speed clock. 
     1.3.1 Coarse Alignment Algorithm 
     Coarse alignment uses the two ASIC ranging registers (the modem transceivers described herein are typically implemented in ASICs), RGSRH and RGSRL. These registers contain information regarding the location of the Barker correlation peak. The registers are each 16 bits, with each bit representing a ½ chip spacing such that the alignment window is +/−8 chips. Referring to FIG. 58, there is shown a diagram of the ranging registers as a function of timing offset. FIG. 58 shows an example of how timing offset affects the values of the RG registers. Note that the 1B and 2A ASICs (two different versions of the ASIC) have a −½ chip offset difference. 
     Coarse alignment tries to align the RGSRH and RGSRL register to equal 0x0000 0x8000 or 0x0000 0xC000 for the 1B ASIC and 0x0001 0x0000 or 0x0001 0x8000 for the 2A ASIC. The coarse alignment algorithm is summarized in the pseudocode algorithm given below (the actual code in the appendices hereto varies somewhat in that it is structured differently and it must handle both versions of the ASIC). 
     Coarse Time Alignment Algorithm Pseudocode Fragment 
     
       
         
           
               
               
             
               
                   
               
             
            
               
                 RXRMR = 1; 
                 % Acquisition mode 
               
               
                 iterations = 0; 
               
            
           
           
               
               
            
               
                 while ((iterations)++ &lt; TIMEOUT)  { 
                 % TIMEOUT = 10 
               
            
           
           
               
               
            
               
                   
                 if (iterations &gt; 4) 
               
            
           
           
               
               
               
            
               
                   
                 fine_shift = 1; 
                 % Gear shift to avoid oscillation 
               
            
           
           
               
               
            
               
                   
                 else 
               
            
           
           
               
               
            
               
                   
                 fine_shift = 2; 
               
            
           
           
               
               
            
               
                   
                 read non-zero value of RGSRH and RGSRL; %try up to 8 times 
               
               
                   
                 if (RGSRH == 0 &amp;&amp; RGSRL == 0) 
               
               
                   
                 return (TIME_ALIGN_ERROR); 
               
               
                   
                 if (peak 1/2 chip to right of center) 
               
            
           
           
               
               
            
               
                   
                 shift_value = fine_shift; 
               
            
           
           
               
               
            
               
                   
                 else if (peak is &gt; 1/2 chip to right of center) 
               
            
           
           
               
               
            
               
                   
                 shift_value = 4; 
               
            
           
           
               
               
            
               
                   
                 else if (peak is 1/2 chip to left of center) 
               
            
           
           
               
               
            
               
                   
                 shift_value = −fine_shift; 
               
            
           
           
               
               
            
               
                   
                 else if (peak is &lt; −1/2 chip to left of center) 
               
            
           
           
               
               
            
               
                   
                 shift_value = −4; 
               
            
           
           
               
               
            
               
                   
                 else return (TIME_ALIGN_DONE) 
               
               
                   
                 SendPowerCommand (shift_value); % Send to RU 
               
            
           
           
               
            
               
                 }% end while loop 
               
               
                 if (iterations == TIMEOUT) return(TIME_ALIGN_ERROR); 
               
               
                   
               
            
           
         
       
     
     The time required for coarse time alignment is dominated by the number of iterations required to converge since each iteration involves the CU having to send a command to the RU via the command and control channels. With a worst-case timing offset of +/−2 chips, the coarse alignment will take at most 10 iterations to converge. With each command taking 64 frames, this amounts to 640 frames or 80 msec. 
     1.3.2 Fine Alignment Algorithm 
     The fine alignment algorithm uses the ClocK Recovery Error Register in FIG. 34 (CKRER) as the metric of timing offset. The CKRER value is derived from the difference of the samples ½ chip on either side of the Barker code correlation peak (“early-late”) shown in FIG.  36 . If timing is perfectly aligned, the samples at  1010  and  1012  in FIG. 36 will equal one another. If timing is off by up to 1 chip, the sign of this metric will indicate the direction of the offset. Since the Barker correlation process is performed prior to chip equalization, the CKRER values are prone not only to noise but also intersymbol interference. As a result of the noise susceptibility, the CKRER values are averaged to mitigate the effects of the noise in this embodiment. 
     The fine time alignment algorithm of the embodiment currently being discussed is given below as a pseudocode fragment. The algorithm consists of reading N CKRER samples and averaging them. This averaged value is used to determine if time alignment had been accomplished. If it is not, the sign of the averaged CKRER value in the CU frame detector of FIG. 34 (the CPU  405  in the CU modem reads the CKRER values repeatedly and averages them) is used by the CU to send a time alignment command to the RU. The process continues until time alignment has been completed or a time-out occurs. 
     Fine Time Alignment Pseudocode Fragment 
     
       
         
           
               
             
               
                   
               
             
            
               
                 RXRMR = 0; 
               
               
                 N = 240; 
               
            
           
           
               
               
            
               
                 while ((iterations) ++ &lt; TIMEOUT)  { 
                 % TIMEOUT = 8 
               
            
           
           
               
               
            
               
                   
                 Read CKRER N times and take abs value of ave =&gt;abs_ave_ckrer; 
               
               
                   
                 /* Check if below threshold or zero crossing */ 
               
               
                   
                 if (abs_ave_ckrer &lt; threshold) || 
               
            
           
           
               
               
            
               
                   
                 (sign(abs_ave_ckrer) == sign(previous_abs_ave_ckrer)) 
               
               
                   
                 return (TIME_ALIGN_DONE); 
               
            
           
           
               
               
            
               
                   
                 else { 
               
            
           
           
               
               
            
               
                   
                 if (abs_ave_ckrer &lt; 0) 
               
            
           
           
               
               
            
               
                   
                 shift_value = −1; 
               
            
           
           
               
               
            
               
                   
                 else 
               
            
           
           
               
               
            
               
                   
                 shift_value = 1; 
               
            
           
           
               
               
            
               
                   
                 }% if 
               
            
           
           
               
            
               
                 } % while 
               
               
                   
               
            
           
         
       
     
     The time required for fine time alignment is dominated by the time required to read the 240 CKRER values. Because of software constraints, 8 CKRER samples are read each 64 frames. So to read 240 CKRER samples each for 8 iterations requires 64*30*8=15360 frames or 1.92 seconds. 
     1.3.3 Time Alignment Phase Shift Compensation 
     Time shifts at the RUs result in phase shifts at the CU. These phase shifts will cause bursts of errors in periodic training and therefore must be compensated. Each high-speed clock (57.344 MHz) offset results in a 22.5 degrees phase offset of the IF- 1  (3.584 MHz) carrier frequency. The RU transmitters transmit at IF- 10  (35.84 MHz), so the IF- 1  carrier is multiplied by a factor of 10. This means that the phase offset is also multiplied by a factor of 10 so that for each high-speed clock offset, the upstream signal is phase shifted by 225 degrees. 
     The precode equalization filters  563  in the RU transmitters can be used to compensate for these changes. By multiplying all four feed-forward coefficients by the negative of the phase shift caused by a timing offset, the phase shift is exactly offset, thereby enacting only the desired time shift of the fine tuning process. 
     1.4 Performance Enhancements and Optimization 
     1.4.1 Coarse Alignment Optimization 
     Worst-case, the coarse alignment algorithm takes 80 msec to achieve alignment. In order to reduce this time, the number of alignment iterations which are required to converge would have to be reduced. Reducing the number of iterations, makes the algorithm more vulnerable to noise. Given that course alignment procedure is effectively only accomplished during initial training, it is not sped up in the preferred embodiment. 
     1.4.2 Fine Alignment Optimization 
     Worst-case, the fine alignment algorithm takes 1.92 seconds to achieve precise alignment. This worst case will likely only be reached in initial training. In periodic training, the system usually performs only one or two iterations for fine alignment. If the CKRER register is read in two iterations (one to detect an offset and another to verify convergence), this gives a fine alignment time of 0.48 seconds. If there are 2000 RU&#39;s in the system, this equates to a minimum training period for each modem of 16 minutes ignoring all of the other aspects of training. This fine alignment execution time can be reduced in some embodiments where such a delay is not acceptable. 
     The simplest option for reducing this time which is implemented in some embodiments is performing less averaging of the CKRER register. In the embodiment implemented by the software appendices appended hereto, 240 CKRER samples are averaged. Preliminary laboratory results have shown that averaging over only 30 CKRER samples yields comparable results. In alternative embodiments where such a reduction is implemented, execution time is reduced by a factor of 8. This gives a worst-case number of 240 msec for initial alignment and a worst-case typical number of 60 msec for periodic alignment. 
     Another embodiment which applies only to 2A modems (and above) uses the demultiplexer memory instead of the CKRER register to sense fine timing alignment status. The demultiplexer memory stores the symbols from all 144 codes output from the demultiplexer. This memory can be used for time alignment by having the RU send only code 4 in BPSK mode. If the RU is perfectly aligned (and ignoring ISI), the CU will see energy only in code 4, with all of the other codes equal to zero. However, if there is a timing misalignment, this will result in the code “spilling” into adjacent codes. By analyzing the direction of spilling, the direction of the timing offset can be determined. 
     In order to make the timing measurements independent of phase (since the phase of the received code is not known), the absolute value of the received codes is taken. However, a blind absolute value will rectify the noise so that it cannot be averaged away. This problem is eliminated by using the soft decisions to take the absolute value. If the RU is aligned within +/− ½ chip, the phase of code 4 will equal the phase of the received symbol. This phase can be used to rectify that transmitted symbol. Since the noise is uncorrelated to the sign of the symbol, it will be reversed in sign. Subsequent averaging of these absolute value symbols will result in an averaging of the noise, but will not average away the transmitted symbols (which is what would happen if averaging was performed before a blind absolute value). 
     Define dmm(3), dmm(4) and dmm(5) as the complex symbols from the demultiplexer memory relating to codes 3, 4, and 5, respectively. Then the time alignment error is given by Equation 6, below, with the absolute value function being performed by multiplying the difference of the adjacent codes by the conjugate of code 4 and then taking the real part of the result. 
     
       
         error=real[( dmm (3)− dmm (5))(( dmm )*(4))]/( abs ( dmm (4))))  (6) 
       
     
     This error is averaged N times and used in exactly the same manner as the averaged CKRER is used in these alternative embodiments as described above in subsection 1.3.2. 
     In order for Equation 6 to be valid, dmm(3), dmm(4) and dmm(5) must all correspond to the same symbol. 
     FIG. 59 shows the preferred structure for the equalizer structure in the RU receivers. Block SE symbolizes the symbol equalizer  1500  in FIG. 30 whereas block CE represents the chip equalizer circuit  764  there. Block Demultiplexer is the demultiplexer  766 , and block R/A is the rotational amplifier in circuit  765  of FIG.  30 . After equalization is achieved in the RU receiver, the coefficients are moved to the local chip equalizer in alternative embodiments represented by FIG.  59 . After equalization is achieved in the symbol equalizer of the CU receiver, the SE coefficients are moved to the precode equalization filter at the RU transmitter. 
     FIG. 60 is a flow diagram of the preferred 2-step initial equalization training algorithm. The purpose of this equalization training algorithm, like the equalization training algorithms disclosed in FIGS. 53A through 53C is to perform equalization training so as to predistort transmissions to minimize the effect on the detection process of phase and amplitude errors induced by channel impairments. In step  1501 , a default value is loaded into SECFF(3) which is a register in the symbol equalizer  1500  in FIG. 30 that stores the coefficient for the main (last) tap of the feed forward equalizer in block  765  of FIG. 30 (the same initial equalization training algorithm applies to both the CU and RU except what is done with the final coefficients differs). Step  1507  trains the main tap by enabling main tap updating for a predetermined number of frames (currently 100 frames in upstream and downstream) and sets the value of an SEKR register in the LMS  830  in FIG. 30 to a value of 866 for upstream equalizations and 666 for downstream equalization training. The SEKR register stores the adaptation coefficient. The rest of the process of FIG. 60 determines when equalization has been achieved by the method of examining the stability of the taps compared to their expected values. When equalization has been completed, it is expected that the main tap will be one and all the other taps will be zero. Step  1509  represents the process of starting to train all the taps by enabling all feedback and feed forward tap updating and setting the SEKR register to 888 for upstream equalizations and 666 for downstream equalizations. Step  1511  represents updating these taps for K 2  frames. K 2  is which is currently set at 20 frames for downstream and 30 frames for upstream equalizations. Test  1513  determines when the equalization process has stabilized. Test  1510  is performed by performing the process of FIG. 61 to determine if the coefficients are close to the expected values for the tap coefficients of 0001. If the equalization has not stabilized, an iteration counter is incremented in step  1517 , the count is checked against the constant M (currently a value of 3) in step  1518 , and if M iterations have not been performed, processing returns to step  1506 . If M iterations have been performed, and the equalization process has not converged, training has failed, and processing proceeds to step  1520  to restart the synchronization process. 
     If the equalization process has stabilized, step  1515  is performed to normalize the symbol equalizer coefficients by dividing them by the value of the main tap SECFF(3). Then step  1519  is performed to convolve the old chip equalizer coefficients and the normalized symbol equalizer coefficients divided by two to derive the new chip equalizer coefficients for downstream or the new precoder coefficients for the upstream. The main tap coefficient of the SE feed forward equalization filter is then set to one and the side tap coefficients of the SE feed forward and decision feedback equalization filters are set to zero for receipt of payload data. Then test  1522  is performed to determine if the equalization process has converged. The equalization convergence test process symbolized by step  1522  is shown in more detail in the flow chart of FIG.  64 . If the convergence has occurred, step  1524  is performed to load the real and imaginary main tap values of SECFF(3) to the rotational amplifier correction routine given below in FIG.  63 . Test  1526  returns processing to step  1506  if convergence has not occurred and passes through the loop are less than 8 for the downstream or less than 5 for the upstream. 
     The process of FIG. 62 is the preferred two-step equalization process which is periodically performed. It is quite similar to the initial equalization process, but fewer iterations are performed. 
     Rotational amplifiers which work on QAM16 constellations can lock improperly on false minima in the error surface of the constellations and cause improper decisions to be made by the slicer. The purpose of the process symbolized by the flow chart of FIG. 63 is to check the rotational amplifier operation in the CU receiver to make sure it has not falsely locked on a local minima. This check is done by comparing the rotational amplifier&#39;s amplitude and phase correction factors against the symbol equalizer main tap correction factor. The rotational amplifier can be considered to be a one tap equalizer which runs all the time. The symbol equalization process is only performed periodically, but it is performed using a QPSK constellation of training data which does not have the false local minima of a QAM16 constellation. Thus, the main tap of the symbol equalizer never falsely locks on a local minima and will always be a correct correction factor to eliminate the effect of phase and amplitude impairments on the channel. The process of FIG. 63 checks for improper locking of the rotational amplifier during reception of each RU&#39;s data by comparing the difference between its amplitude and phase correction factors to the amplitude and phase correction factors of the main tap of the symbol equalizer. If the difference is too large, the rotational amplifier has falsely locked, and it must be corrected by setting its correction factors to the amplitude and phase correction factors of the main tap of the symbol equalizer. Step  1530  symbolizes the step of setting the square of the amplitude, Amp racm , of the rotational amplifier correction factor Amp racm e jø  to the sum of the squares of the RU correction factor stored in memory  796  in FIG. 31 for the particular RU whose data is being received. Step  1532  then calculates the square of the amplitude correction factor for the main tap of the symbol equalizer by setting it equal to the sum of the squares of the real and imaginary parts of the SE main tap coefficient, i.e., SECFFl 3  and SECFFQ 3 . Step  1534  then calculates the phase difference between the rotational amplifier correction factor and the symbol equalizer main tap by calculating: 
     Phase dif =l racm  SECFFQ 3 −SECFFl 3  Q racm . Test  1536  then determines whether the absolute value of the difference between the rotational amplifier correction factor and the SE main tap amplifier correction factor is less than an amplitude threshold. If it is not less than this threshold, the rotational amplifier has falsely locked, and processing proceeds to step  1538  to correct the situation by loading the SE main tap correction factor into the memory for the rotational amplifier as the new correction factor for this RU. Test  1540  makes a similar comparison for the phase difference between the rotational amplifier and the symbol equalizer main tap. If the phase difference is too large, processing proceeds to step  1538  again. In other words if either the amplitude difference or the phase difference between the correction factors of the rotational amplifier and the main tap of the SE is too large, the rotational amplifier correction factor for that RU is set equal to the SE main tap value. If both tests are passed, step  1542  symbolizes the process of not making any correction to the rotational amplifier correction factor. 
     FIG. 64 is a flow chart symbolizing the process of step  1522  in FIG. 60 in determining whether the equalization training process has converged. Generally, if the equalization training process properly converged, the SE main tap (tap  3 ) correction factor will be one and the SE side taps (taps  0 - 2 ) will be zero. The process of FIG. 64 determines whether the ratio of the amplitude correction factor of the SE side taps to the amplitude correction factor of the SE main tap is smaller than a threshold. If it is, then the equalization training process has converged. If not, the equalization training process has not converged. Step  1544  calculates the amplitude of the side taps of the SE as the summation of the sum of the squares of the real and imaginary components of side taps  0  through  2  of the FFE filter in circuit  765  of FIG. 31 plus the sum of the squares of the real and imaginary parts of the side taps  0  through  3  of the feedback DFE filter  820  in FIG.  31 . Step  1546  calculates the amplitude correction factor of the SE main tap as the sum of the squares of the real and imaginary parts of the SE main tap correction factor. Step  1548  calculates the ratio of the SE side tap to main tap amplitude correction factors, and step  1550  compares this ratio to the threshold of convergence which can be experimentally determined. If the ratio is not less than the threshold, the equalization process has not converged, as symbolized by step. If the ratio is less than the threshold, the equalization process has converged, as symbolized by step  1554 . 
     Referring to FIG. 61, the details of the process represented by step  1510  of FIG. 60 in determining whether the equalization process has stabilized are shown. When the equalization training has converged properly, the SE side taps will be zero or small and the SE main tap is expected to be near one. The basic test performed in FIG. 61 is to compare the ratio of the amplitude correction of the SE side taps to the amplitude correction of the SE main tap to make sure the ratio is below a predetermined threshold. If it is, then equalization has converged. Other subtests also exist. Steps  1560  and  1562  represents one subtest to determine if each of the SE&#39;s FFE side taps  0  through  2  are smaller than a predetermine threshold Thrld coef . If any one of the feed forward side tap coefficients is not smaller than the threshold, processing proceeds to step  1564  (representing steps  1516  and  1518  in FIG. 60) to declare the equalization unstable and return to step  1506  in FIG. 60 to begin the all taps training again. Steps  1566 ,  1568  and  1570  represent a similar subtest for the side taps of the SE feedback side taps  0  through  3  (SECFB k ). If any one of these feedback filter side taps is larger than the threshold, the equalization training process will be declared unstable. Step  1570  calculates the composite amplitude of the side tap correction factors for both the feed forward (FFE filter in circuit  765  in FIG. 31) and feed back (DFE filter  820  in FIG. 31) SE filters as the summation for taps  0  through  2  of the squares of the real and imaginary components of each tap&#39;s coefficient for the FFE filter, plus the summation for taps  0  through  3  of the squares of the real and imaginary components of each tap&#39;s coefficient for the DFE filter. This sum is called Amp side . Step  1572  calculates the amplitude of the SE filter FFE main tap coefficient as the sum of the squares of the real and imaginary parts thereof, and assigns this sum to variable Amp main . Step  1574  calculates the ratio Amp side /Amp main , and step  1576  compares this ratio to a threshold of stability Thrld stable . The ratio is expected to be small for a stabilized equalization process, so step  1578  (representing a vector to step  1512  in FIG. 60) is reached if the ratio is less than the threshold, meaning that the equalization process is equalized. Step  1580  (representing steps  1516  and  1518  in FIG. 60) is reached if the ratio is greater than the threshold, meaning that equalization process has not stabilized. 
     Power Alignment Procedures 
     The near-far problem in upstream transmission is solved partially by the ranging process which does a coarse power alignment. In the preferred embodiment, a fine tuning of the power level of each RU is also performed so that the power of each RU&#39;s transmission as received at the CU is approximately the same. The power level of each RU at the CU is detected by a gain detector which, in the preferred embodiment, if the main tap amplitude correction factor of the SE. The RU transmitter power of each RU is adjusted with the help of information from the CU gain detector. The CU expects specific received power levels from each RU (0 dBmv). The RU transmitter power ranges from 32 dBmv to 52 dBmv. The power alignment fine tuning process is accomplished as follows and as depicted in the flow chart of FIG.  65 . 
     1) CU asks RU to transmit all training codes. These are the 8 training codes used for the equalization training process. 
     2 ) CU run the SE equalization process for N frames (N=80) (when the training codes are being sent, the CPU in the CU enables the SE in the CU receiver to iterate for 80 frames) with 
     SEKR=0x0855H (SEKR is a register in the LMS  830  in FIG. 31 which stores the adaptation coefficient—in this case the adaptation coefficient is set equal to this constant in hex notation, as symbolized by step  1600 —this adaptation coefficient is selected to insure rapid convergence during power alignment which is important in huge systmes with many RUs although slower convergence can be selected in smaller systems) 
     SECFF(3) enabled and all other taps disabled (only the main tap of the FFE filter in the SE is enabled for fine tuning of the power alignment so as to act as the gain detector, and all side taps are disabled, as symbolized by step  1602 ). 
     3) Calculate delta 1 =(SECFF(3){circumflex over ( )}2-1FFFH)/k 1 , where k 1 =64 (this is the process of calculating the amplitude of the coefficient of the FFE filter main tap in the SE as the sum of the squares of the real and imaginary components I and Q; this amplitude has subtracted from it the expected value of the expected main tap value, 1FFF hex, when power alignment has been achieved; the difference is then divided by the constant  64 , which in this embodiment is set equal to 64; the result is called delta 1 —all as symbolized by step  1604 ). The power alignment process of FIG. 65 differs from the power alignment process of steps  1108  through  1112  of FIG. 53A in that the process of FIG. 65 uses the main tap of the SE filter as the gain sensor in the CU after convergence by the SE whereas in the process of FIG. 53A, the gain control number for power alignment is derived by convergence of an adaptive gain control circuit comprising the slicer, slicer error signal, control loop  781  and the variable gain amplifier  788  in CU receiver. Further, there is no necessity in the power alignment process of FIG. 65 to set the RU gain level at one before transmitting the training data. 
     4) If abs(delta 1 )&lt;TH, Power alignment is done (this is the process of comparing delta 1  to a small threshold; if delta 1  is below the threshold, power alignment is completed and processing vectors to step  1608 ; if not, a message is sent to the RU to lower its power in the next step—all as symbolized by step  1606 ) 
     5) If not, send to RU a power adjustment factor equal to delta 2 =delta 1 *2{circumflex over ( )}(−ki). The factor delta 2  is a power adjustment factor telling the RU a correction factor by which to adjust its power, with the correction factor larger for larger differences over the threshold than for smaller differences; the correction factor is equal to delta 1  times a constant—this process is symbolized by step  1610  in FIG. 65) 
     6) RU updates TXLVLR=TXLVLR+delta 2  (this is the process symbolized by step  1612  wherein the RU updates the value in register TXLVLR which controls the power level of the RU transmissions by adding the value of delta 2  to the register contents). 
     Steps  1612 ,  1614 ,  1616  and  1618  in FIG. 65 are the steps which determine if the number of desired iterations of the power alignment process have been achieved. 
     Boundless Ranging Preferred Embodiment Previous Ranging Scheme 
     With the exception of the boundless ranging scheme wherein the CU calculates the total turnaround time for each RU and sends that data to the RU, the previously described ranging schemes do not deal with boundless distance ranging. Boundless ranging schemes are especially useful in large systems where there is an RU at the head-end in addition to many RU modems distributed throughout a system coupled by multiple fiber nodes, a situation depicted in FIG.  66 . If the RUs beyond optical nodes  1620  and  1622  have more than one frame offset in TTA and are aligned to different gaps than the RU  1624  at the CU, the boundless ranging problem of possible confusion of which codes to use in the CU in decoding particular frames from the RUs exists. 
     In addition, the previously described ranging scheme requires a calibration procedure with the modem at the optical node in order to find the latency from the head-end to the optical node. The other ranging schemes described herein (with the exception of the alternative boundless ranging scheme mentioned briefly above, required that all RUs be close enough (16 kilometers) of the CU such that their TTA was less than one frame (125 microseconds). In these embodiments, it was assumed that there was an RU at the optical node and the CU receive window and gap was offset from the CU transmit gap by the amount of the TTA to the first optical node such that if the RU at the optical node sent back the CU barker code immediately without delay, it would arrive at the CU later by the TTA to the optical node. This required a calibration procedure to determine the TTA to the optical node so as to offset the CU receive window properly. This was inconvenient, but is unnecessary in the boundless ranging embodiment disclosed here. 
     In addition, the previously described ranging schemes uses a sequence of 8 pulses during all the ranging stages which adds a lot of interference to the data portion, and requires a longer time. In other words, the old ranging scheme had the RUs continuously sending their ranging IDs which had 8 barker codes. If these 8 barker codes were misaligned, they landed on payload data and could cause errors. The boundless ranging scheme described here uses only one barker code transmission at a time until confirmation is received from the CU that it landed in the gap. 
     The preferred boundless ranging process has all RUs adjusting their delays to hit the gap after the frame number from receipt of the Barker code from the CU which the farthest away RU can hit (currently the 13th frame for a system which spans 100 miles). Each RU hits the same gap using an offset number plus a transmit frame timing delay value T d . The offset number is the total turnaround time (TTA) from the CU to that RU and back in frames. Any fraction of the TTA that is less than a complete frame is the value of T d . Note that it is only necessary for all the RUs to align to the same gap and keep track of CU frame numbers where the codes are being time shared. In embodiments where the codes are not time shared such as where each RU always transmits on the same code or codes, it is only required that the frame boundaries be aligned, and this is only necessary in order to minimize ISI. If other methods of limiting, eliminating or compensating for ISI are available, even this frame synchronization can be eliminated and regular CDMA used in the upstream channel. 
     This can best be understood by joint reference to FIGS. 66 and 67. FIG. 67 shows how the offset number is used to achieve frame synchronization. Suppose RU  1626  is 100 miles away from CU, and that, assuming that RU  1626  sends its ranging signal immediately upon receiving the Barker code from the CU, that ranging signal does not arrive until the gap following the 13th frame from the gap in which the CU originally transmitted its Barker code. For boundless ranging RU  1624  has to hit the same gap and puts its data in the assigned frame. 
     Proper accounting for assigned frames is the reason for boundless ranging. Since there are only 128 channels, but there may be 2000 RUs, the CU controls the situation as follows. RUs having data to transmit, send management and control messages to the CU saying they need bandwidth. The CU assigns one or more codes to the RUs according to whatever bandwidth allocation scheme is in use. 
     ***Dynamic bandwidth allocation allows as many 64 kbps streams or channels as necessary to be allocated to a particular service so that high demand applications such as video teleconferencing or high speed internet access can be supported simultaneously with low demand applications like telephony over the same HFC link. Bandwidth allocation is managed at the CU through an activity status table in each RU and the CU that indicates the status of each timeslot and code assignments. The CU updates the RU tables by downstream messages. Bandwidth can be guaranteed upon request while other services with more bursty traffic may contend for the remainder of the total 10 Mbps payload. 
     The bandwidth assignments are sent downstream as code numbers to use during specific frame numbers. Referring to FIG. 67, each RU receiver section  1630  includes a frame counter in the frame detector that increments each time a Barker code is received from the CU. That information plus the kiloframe markers in the pilot channel data, tell the RUs which CU frame they just received. 
     Suppose RU  1624  in FIG. 66 was assigned code 1 for use during frame  1000 , and RU  1626  was assigned code 2 for use during frame  1000 . For frame synchronization to exist, RU  1626  must transmit its frame  1000  using code 2 at a time 13 frames earlier than RU  1624  transmits its frame  1000  using code 1. This is accomplished by using an offset number. Basically, the offset register  1632  in FIG. 67 is set to −13 in RU  1624  and is set to 0 in RU  1626 . The value in the offset register is subtracted by subtractor  1634  from the CU frame count determined by the receiver  1630 . The result is the frame count that the RU transmitter section  1636  uses to control when it transmits its frame  1000 . Thus, the transmitter  1636  of RU  1626  100 miles away from the CU reaches frame count  1000  13 frames earlier than the transmitter  1636  of RU  1624  which is at the CU. 
     Therefore, the RU  1626  transmits its frame  1000  using code 2 13 frames earlier than RU  1624  transmits its frame  1000 . As a result, frame  1000  from each of the RUs  1626  and  1624  reach the CU at the same time (CU frame count  1000 ) and are properly demultiplexed using the codes assigned to these RUs for frame  1000 . 
     There follows a detailed discussion of a ranging process which supports boundless ranging. 
     The Preferred Ranging Algorithm For Boundless Ranging 
     The ranging algorithm is the procedure by which an RU aligns its coarse transmission frame timing so that its frame start (i.e. the beginning of the frame gap) will be received by the CU exactly when the CU begins a new frame. This will cause all the RU transmitted frames to be received in the CU aligned to each other and to the CU. 
     The ranging process is performed after the RU is powered on and finishes its downstream initialization (clock and carrier recovery) functions and its received frames are aligned to its clock. This sequence of events allows the RU to be able to receive the data sent in the downstream by the CU. 
     Ranging is the first step of the upstream initial training process which includes the following steps: 
     1. Ranging 
     2. Time alignment. 
     3. Power alignment. 
     4. Time alignment. 
     5. Power alignment. 
     6. Equalization. 
     7. Repeat steps 4 through 7 N times. 
     The ranging process should: 
     1. Align the RU&#39;s frames to within +/−1 chip of the exact timing required to achieve frame synchronization 
     2. Align the RU&#39;s power level to achieve detectable power levels but not exceed the power level that increases the bit error rate of the operating RUs. 
     The objective of the ranging process are as follows. 
     1. To find the delay between the instant that the RU under consideration receives the frame start from the CU (in the downstream channel) until it has to begin transmitting the frame start in the upstream so that it will be received by the CU aligned with the CU frame start. FIG. 68 represents the frame start propagation delays along the downstream channels and the required delays Δ 1 , Δ 2  for each RU to hit a gap assigned to the group of RUs to which that RU is assigned. 
     2. To find the power level with which the RU has to transmit its ranging signals, so that the CU receiver will be able to detect it, but at the same time will not introduce additional noise in the running RUs data channels (in the upstream). That is, it has to find the minimum power level that the RU should transmit the ranging signals, so that the CU receiver will detect them with high enough detection probability without unduly interfering with data being transmitted by other RUs should the RU ranging signals arrive at the CU mistimed and in the middle of another RU&#39;s payload data. 
     3. To find the frames offset that the RU under consideration has to have in its frame counter in order to be able to align its transmitted frames indices to the received frames in the CU. This offset is required to synchronize all the RUs in the frame level for control purposes (such as time-slot allocation, retraining initialization, etc.) and is needed because of the channel total-turn-around time (TTA) due to the required maximum distance between the CU and the farthest RU (100 miles). This element of the ranging is also known as the “boundless distance ranging”. FIG. 69 shows a channel with TTA of 3 frames. 
     The main idea in the ranging process is that the RU performs an efficient search of the [delay, power] plane in order to find the appropriate delay and power that will satisfy objectives 1 and 2 above. The RU has to transmit the ranging signal so that it will be received by the CU&#39;s receiver at a window located at the center of its gap as shown in FIG.  70 . The window in the center of the gap during which ranging signals are received in this embodiment is shown at  1640  and is 6 chips wide. The gap, of which gaps  1642 ,  1644  and  1646  are examples, is located at the beginning of every frame, and serves as a listening window for the CU receiver. The CU serves as a sensor for RU ranging signals that provides feedback signals when detecting activity in the gap. In order that the CU&#39;s receiver will receive the ranging signal of some RU, the signal must be received within the CU gap (and more specifically, in the ranging listening window  1640 ) and with high enough power. 
     The ranging algorithm described here allows many RUs to perform their ranging at the same time and still achieve the separate frame synchronization of each of the RUs. Each RU sends a ranging signal that is a series of 17 pulses (one pulse per frame—gap). These 17 pulses include a starting pulse and 16 ranging ID pulses out of which 8 are “0” and 8 are “1”. A “1” signal is represented by the presence of a Barker coded pulse with length of 13 chips. A “0” is represented by the absence of transmission of the Barker code. The ranging ID is randomly picked by each RU. It takes 17 frames for an RU to send its ranging signal. Because the maximum distance between the CU and the farthest RU is 100 miles, the TTA time is 16 frames. This means that the possible received ranging ID will be located within 32 (=17+16−1) frames in the CU. 
     The CU receiver looks for Barker signals in the six middle chips (i.e. the ranging listening window  1640 ) of the gaps of 32 successive frames whose positions (indices) were predefined by the CU in the ranging solicitation message it transmitted. The CU translates these 32, 6-chips-long vectors, into 6 32-digit vectors V 1 , V 2  . . . V 6  where V 1  denotes all the received values (0/1) of chips number  1  in the successive frame listening windows  1640 , and V 2  contains the same for chip number  2  in the ranging listening windows, and so on. In the boundless ranging embodiment described below, the listening window in the middle of each gap is 8 chips wide, so vectors V 1  through V 8  are built by the CU CPU in the same fashion as described above. 
     FIG. 71 is a pictorial description of the 6 chip listening window translation or mapping of the contents to the 6 chips of the listening window of 32 consecutive frames into the vectors V 1  through V 6 . In each of these six vectors, the CU looks for the structure of a ranging signal (17 pulses with 1 start bit and in the next 16 cell positions, the CU looks for the structure of a ranging ID—8 ones and 8 zeros. When the CU identifies this structure in some Vk vector it defines it as a “valid ID”. 
     In general, the CU receiver can have the following possible cases for each vector: 
     1. The CU does not detect any signal in these vectors—in this case it sends an “empty” message in the downstream. 
     2. The CU detects only valid ID (up to six)—in this case it will send the “valid ID” message in the downstream with the detected IDs list. 
     3. The CU detects signals that do not have the structure of a valid ID—in this case the CU treats the received signal as being the result of contention of at least two RUs and sends a “collision” message. 
     Since the CU examines six vectors, it can also have the mixed cases: “valid+empty”, or “valid+collision”. 
     The scan of the delay-power plane is performed so that the RU scans all the relevant delays for a given power, and if this does not get the CU&#39;s response, the RU increases its power by a given step (ΔP), and re-scans all the delays. Assume A is the initial scanning point (minimum delay and minimum power). From A, the scan goes with constant power and increasing delays until the maximum delay is checked. If this is not sufficient, the power is increased and the delay is scanned from minimum to maximum values. Once the RU Barker code transmission has the correct delay and power level values to be received by the CU receiver in the gap, the CU will send an appropriate message that will stop the scan. If the message is “valid ID” with the RU&#39;s ranging ID, the RU completes its ranging while performing a last update of its delay according to the CU&#39;s adjustment data that is sent to the RU. 
     If the CU gets a non-valid ID with at least three “ones”, it responds with a “collision” message that puts all the RUs that are ranging in this instant into a contention resolution mode. The contention resolution is performed in the ranging algorithm via a binary tree algorithm to “flip the coin” as previously described. 
     Summary of Steps of Preferred Boundless Ranging Process 
     (1) CU solicits for ranging and scans a number of following frames equal to or greater than the TTA in frames to the farthest RU. 
     (2) RU transmits Barkers continuously with 4 chip delay increase. After each iteration through all possible delays, the power increases by k dB (k is a constant which can be determined experimentally). 
     (3) CU sends “ACTIVITY DETECTED IN FRAME#”. 
     (4) RU does not know which value of T d  resulted in hitting the gap or even if it was the RU that hit the gap. There is a 16 frame ambiguity or 16 different possible delay values that could have caused the gap to be hit for the farthest RU in a system with a 16 frame TTA span. As a result, it assumes it was the RU that hit the gap and goes back to the delay value it used for the frame number 16 frames ago (for a system with a TTA span of 16 frames) and starts negotiation with the CU by sending one Barker code at a time and waiting for a reply. That is, a value for T d  will be picked starting with the T d  used 16 frames ago and a Barker code will be sent. The RU will then wait 16 frames for a reply message from the CU regarding whether activity was detected in the gap. If no such message is received, the next value for T d  will be selected, and another Barker code will be sent with the RU again waiting 16 frames for a reply. 
     This is one Barker code at a time scheme is faster than some of the alternative ranging schemes previously described wherein the RU sends its entire ranging ID each time it picked a new delay value since in those alternative schemes a number of frames equal to the number of bits in the ID was consumed for each value of T d  before the RU knew whether that was the correct T d  value. 
     (5) CU sends “ACTIVITY DETECTED IN FRAME#, START CONTENTION RESOLUTION”. 
     (6) all RUs that are ranging, transmit an ID sequence comprised of a start bit (always a logic 1) and 8 random Barkers out of 16 in the 16 consecutive frames following the start bit (in some embodiments, the ID code for each RU can be fixed and will be comprised of a start bit and an even number of consecutive bits exactly half of which are “1s”—in some embodiments, the ID code may have some other known number of 1&#39;s which are more or less than half with contentions being detected by detection in one vector of a number of 1s in excess of a predetermined threshold). 
     (7) CU looks for valid ID (exactly 8 of the 16 gaps following the start bit contain barker codes), and, if a valid ID is found, broadcasts a message containing the VALID ID (the actual ID found), the FRAME# (frame # of the frame in which the start bit arrived of the valid ID), and a CHIP OFFSET (instructions on which way to adjust the value of T d  to start the fine tuning process). 
     (8) each RU with an ID found by the CU recognizes its ID in the CU broadcast message or messages, calculates its offset value and does fine tuning (one RU at a time under control of the CU) to center its barker code exactly in the middle of the gap. 
     (9) For centering and confirmation, CU sends a downstream message containing PREVIOUS ID (the ID it just received) and a request, TX ANOTHER ID. 
     (10) RU sends another random ID (randomly selected and usually different from the first ID it used in the ranging process). CU broadcasts new IDs found and RU knows it has completed ranging unless contentions found. 
     (11) If there are contentions, CU sends “CONTENTIONS DETECTED”. 
     (12) each RUs that is ranging starts binary tree algorithm for contention resolution as previously described, and some stop ranging and some continue ranging. 
     More Details on Preferred Boundless Ranging Process 
     The preferred boundless ranging process is broken generally dowin into several phases: Activity detection; Contention detection resolution, and authentication, and Frame alignment and gap centering. 
     In the preferred embodiment, only one pulse is transmitted per frame by each RU which is ranging for purposes of activity detection. For contention detection and resolution and frame alignment, gap centering nand authentication, the RUs send a 17 bit ID which comprises a sequence of one start Barker code followed by an ID comprised of 16 ON or OFF “bits” of which precisely 8 will be ON. 
     The details of the preferred boundless ranging process for use in an SCDMA embodiment where the codes are time shared are as follows. 
     (1) the CU solicits for ranging continuously and then analyzes each X frames following each Barker code transmission for activity. X is equal to the TTA to the farthest RU in number of frames, so if the farthest RU is 16 frames out, the 32 frames following each Barker code transmission from the CU will be analyzed for activity. 
     (2) each RU which is ranging transmits Barkers in consecutive frame gaps with a 4 chip delay increase in T d  until CU reports activity detection. RU starts with a small power level so as to not cause excessive interference if it is not properly frame synchronized at the current value for T d . If it does not receive feedback from the CU at this power level after scanning all the possible delays, the RU increases the power by k 3 =3 dB and starts scanning again with 4 chip delay increases during each successive transmission. The power for ranging is limited to maximum power allowed less 4 dB. At this time the RU is only trying to hit the nearest gap and does not care what its offset is. That offset will be deduced later from a message from the CU. 
     In other words, in the beginning of the ranging process, the RU transmitter FRAME # has 0 offset relative to the receiver FRAME #, and the CU FRAME # is the same in the transmitter and the receiver. 
     (3) Activity detection: As CU detects activity in the gap, it notifies the RUs “activity detected in frame #m”. The FRAME# is sent to the RUs to eliminate any software delay. No RU is allowed to start new ranging processing after the first activity detection. The RU optionally increases the power by k 4 =1 dB to ensure better Barker detection. 
     The TTA for the RU or RUs that hit the gap is unknown at this point. Therefore the RU assumes that it is the RU that hit the gap and needs to determine which of the values for T d  it used for previous Barker code transmissions which caused it to hit the gap. The RU therefore backs up its delay value and starts a one Barker code at a time negotiation process with the CU to attempt to locate the value for T d  which caused it to hit the nearest gap. The value of T d  is always less than one frame or 125 microseconds. For RU located up to 100 miles from the CU the maximum latency or TTA delay is 13 frames obtained from:          TTA   max     =         2   ×   100                 miles       0.65   ×   3   ×     10   8                   m        /        s       =       1.6                 ms     =     13                 frames                         
     For margin, assume a maximum TTA of 16 frames to the farthest RU. The RU uses the same chip delay used in frame #m identified in the CU&#39;s “activity detected” message in order to transmit the first Barker code. Only one Barker code is transmitted, and a reply from the CU is awaited for k 5  frames (an experimentally determined number of frames). If the Barker is not detected, the RU continues to try all the 16 possible delays by increasing the delay by 4 chips for each trial. Each trial takes k 5  frames until the RU receives feedback from the CU that activity has been detected in the gap. The value of k 5  is set according to due to TTA and software delay. In alternative embodiments, a few pulses can be together. 
     Once the CU detects a pulse inside the gap, it sends to the RUs a message indicating “a pulse was detected in frame #n”. This frame number can be used to set the offset value in some embodiments where there is known to be no contention (such as embodiments where only one RU is allowed to range at a time), but, in the preferred embodiment, since the RU is not yet sure it is the RU that hit the gap and there is only one RU in the gap, the offset number cannot yet be calculated. 
     (4) Contention detection and resolution: After the pulse was detected inside the gap, there are a few possibilities. First, there could be one pulse inside the gap was detected as well as other pulses which landed in the data portion. The source of the pulse in the gap can be from one RU, a contention of two or more RUs, or a noise hit. Second, there could be more than one pulse inside a gap or more than one pulse in several gaps. To resolve any contentions between multiple RUs, a contention resolution scheme similar to the scheme previously described herein can be used but with the addition of a start bit to accomodate the fact in boundless ranging that there is no longer any limitation that TTA max  be limited to one frame. Thus, the CU sends a downstream message, “ACTIVITY DETECTED IN FRAME #XX, START CONTENTION RESOLUTION”. 
     (5) ***To determine if there are any contentions in the preferred boundless ranging embodiment, the RUs transmit a sequence of 17 bits comprises of a start bit (a Barker code) and 81&#39;s out of 16 successive gaps where a “1” is a gap with a Barker code transmission in it. Vectors V 1  through V 8  are created by the CU in the same manner described above for creation of vectors V 1  through V 6  for the 6 chip listening window embodiment. Contentions are detected by the CU when the number of 1&#39;s in any particular chip vector V 1  through V 8  is more than 3 and different from 8. If there are contentions, the CU goes into contention resolution phase. The CU looks for a valid ID sequence in each vector in the process of examining each vector for contentions. A valid ID is found when exactly 8 of 16 gaps following the start bit having a “1”, i.e., a Barker code in them during an interval of 17+16=33 frames (17 frames for the start bit and 16 bit ID and 16 frames for TTA involving propagation of the command START CONTENTION RESOLUTION to the farthest RU in the system and propagation of the start bit back to the CU from the farthest RU. Each valid ID sequence starts with a start bit, and it is the frame number during which the start bit arrived at the CU which is the frame number in which the valid ID is deemed to have arrived if a valid ID is found. The use of the start bit in this manner give definiteness to the offset calculation by insuring that each valid ID starts with a 1 so that its time of arrival can be determined with no ambiguity. The receipt of the valid ID insures that there are no contention, because if more than one RU is in the gap, more than 8 “1s” will be received in the 16 gaps following the start bit. When the valid ID is received, it is broadcast downstream in a message “VALID ID, FRAME #, CHIP OFFSET”, and the RU which transmitted the ID will know that it is the RU that hit the gap by virtue of seeing its ID in the downstream message. 
     An example of a table obtained at the CU in looking for valid IDs is given in FIG.  72 . Each row in the table represents one of the vectors V 1  through V 8 . Note that in this particular embodiment, the listening window for Barker codes is 8 chips in width. In alternative embodiments such as the embodiment described next above, the listening window can be 6 or even 4 chips in width. 
     The table of FIG. 72 shows 8 valid IDs received during the window, one during each of the 8 chips of the listening window. The start bit for each valid ID is shown in a shaded box, and is used to detect the beginning of the sequence. Using this scheme the CU can detect collisions and ranging IDs for up to 8 different RUs. Contentions are detected on a per chip, i.e., a per vector basis in the 8 chip listening window in each gap. 
     (6) After a valid ID is detected, the CU broadcasts messages for each correct ID. Each message is comprised of: valid ID (the actual ID received), FRAME number of the frame in which the start bit of the valid ID was received, and the number of chips the start bit was offset from the center of the listening window. The RUs which see their ID in the downstream messages know they have hit the gap and that there is no contention on the particular chip in the listening window in which they have landed. Each RU which recognizes its ID in the downstream message “VALID ID xxxxxxxxx, FRAME #, CHIP OFFSET” then calculates its offset number by using the CU frame number contained in this downstream message containing that RU&#39;s valid ID (this is the CU receive frame number during which the start bit of that RU&#39;s valid ID was received) and the CU transmit frame number count from the CU transmit frame counter in the frame detector of that RU&#39;s receiver (which matches the CU transmit frame count) contained in the message from the CU saying “ACTIVITY DETECTED IN FRAME #, START CONTENTION RESOLUTION (Send your ID)”. The difference between these two frame numbers is the propagation time in whole frames of a signal from the CU to the RU and back to the CU which is equal to the TTA in whole frame for that RU. That offset number is set into the offset register  1632  in FIG. 67 so as to achieve frame synchronization in this boundless ranging process. 
     In some embodiments, the Barker code listening window inside the gap is only 4 chips wide instead of 8, to prevent any of the Barker energy from being spilled into the data portion. Note that contentions are determined on a per chip basis in the listening window, so as long as only one RU has its Barker codes landing on that particular chip in every gap listening window, there is no contention on that chip. To avoid complication of the centering or fine tuning process, the CU in the preferred embodiment will fine tune only one RU at any particular time by messages in the downstream command and control channels telling the RUs which is to fine tune at any particular time. 
     (7) Fine Tuning: The RUs that see their valid ID in the downstream messages, then employ frame alignment and Barker centering by correcting the delay in terms of frames and chips to complete their ranging when so instructed by the CU. 
     (8) Contention resolution: When the RUs get a message of contention from the CU instead of the message “VALID ID xxxxxxxx, FRAME #, CHIP OFFSET”, the RUs “flip the coin” using a binary tree algorithm, to decide whether they continue ranging. Each RU in contention resolution mode has a probability of ½ that it will retransmit the 17 bit ID sequence. 
     The CU should have the following commands for contention resolution: 
     (A) “retransmit the sequence with probability of x, x=½ or 1” 
     (B) “retransmit the sequence with probability of x, x=½ or 1 only if the RU transmitted the sequence one stage before” 
     (C) “retransmit the sequence with probability of x, x=½ or 1 only if the RU transmitted the sequence one or two stages before” 
     (9) Centering and confirmation: The CU asks each successful RU which has centered its ID to transmit another random ID at the gap center in order to reduce the probability of errors. 
     After the CU finishes with sending IDs and contention resolution it should notify all the RUs that they are allowed to start ranging or restart ranging. The RUs that restart ranging should continue their ranging from the state it was stopped. 
     Although the teachings of the invention have been illustrated herein in terms of a few preferred and alternative embodiments, those skilled in the art will appreciate numerous modifications, improvement and substitutions that will serve the same functions without departing from the true spirit and scope of the appended claims. All such modifications, improvement and substitutions are intended to be included within the scope of the claims appended hereto.