Patent Publication Number: US-2021186553-A1

Title: Medical device

Description:
The present invention relates to the field of medical devices and in particular, although not exclusively, to medical cauterization and cutting devices. The invention also relates to drive circuits and methods for driving such medical devices. 
     Many surgical procedures require cutting or ligating blood vessels or other internal tissue and many procedures are performed using minimally invasive techniques with a hand-held cauterization device to perform the cutting or ligating. Some existing hand-held cauterization devices use an ultrasonic transducer in the cauterization device to apply ultrasonic energy to the tissue to be cut or ligated. Other hand-held cauterization devices apply RF energy directly to the tissue/vessel being cauterized via forceps of the device. 
     The present invention aims to provide an alterative surgical device that is able to apply ultrasonic energy or RF energy to the vessel or tissue to be cauterized. Other aspects of the invention relate to the way in which control circuitry is provided to select between the different operating modes. Other aspects of the invention relate to the way in which voltage and current measurements can be made in the circuit design for reporting to a controller, such as a microprocessor; and to the way in which control circuitry can be provided to ensure that too much current is not drawn from the battery. 
     According to one aspect, the present invention provides a medical device comprising: an end effector for gripping a vessel/tissue; an ultrasonic transducer coupled to the end effector; a drive circuit coupled to the end effector and to the ultrasonic transducer and operable to generate a periodic drive signal and to provide the drive signal either to the ultrasonic transducer or to the end effector; and a controller operable to control the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector. 
     In one embodiment, the drive circuit comprises a first resonant circuit having a first resonant frequency and a second resonant circuit having a second resonant frequency that is different to the first resonant frequency, wherein the first resonant frequency corresponds to a resonant characteristic of the ultrasonic transducer and wherein the controller is operable to control the drive circuit so that the drive circuit generates a drive signal having a frequency corresponding to the first resonant frequency when the drive signal is to be applied to the ultrasonic transducer and so that the drive circuit generates a drive signal having a frequency corresponding to the second resonant frequency when the drive signal is to be applied to the end effector. 
     A signal generator may also be provided that is coupled between the controller and the drive circuit for generating a cyclically varying voltage from a DC voltage supply in dependence upon control signals from the controller and for supplying the cyclically varying voltage to the first and second resonant circuits of the drive circuit. 
     The controller may be arranged to vary the period of the drive signal about the first resonant frequency or the second resonant frequency to vary the energy supplied to the vessel or tissue gripped by the end effector. The controller may vary the period of the drive signal so that the frequency of the drive signal varies around the first resonant frequency within 0.1% to 1% of the first resonant frequency; or so that the frequency of the drive signal varies around the second resonant frequency within 40% to 60% of the second resonant frequency. 
     Typically, the resonant characteristics of the first and second resonant circuits vary with the tissue or vessel gripped by the forceps and in one embodiment, the controller is configured to vary the period of the drive signal to track changes in the respective resonant characteristic. 
     An ultrasonic waveguide may be provided that is coupled to the ultrasonic transducer for guiding ultrasonic energy generated by the ultrasonic transducer towards the end effector. The end effector may comprise first and second jaws and the second resonant circuit may be electrically coupled to the first and second jaws of the end effector. For example, the first jaw of the end effector may be electrically coupled to the waveguide and the second resonant circuit may be electrically coupled to the first jaw of the end effector via the ultrasonic waveguide. In some embodiments, the first resonant circuit is electrically coupled to the ultrasonic transducer and to the waveguide. 
     Sensing circuitry may be provided for sensing a drive signal applied to the ultrasonic transducer or to the end effector. In one embodiment, one or both of the first and second resonant circuits may comprise at least one of an inductor coil, a capacitor and a resistor and wherein the sensing circuitry may comprise an op-amp circuit for sensing the voltage across the inductor coil or the capacitor or the resistor and for converting the sensed voltage to a sensor signal suitable for inputting to the controller. In an alternative embodiment, one or both of the first and second resonant circuits may comprise an impedance element that is connected between the resonant circuit and a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure. In this case, the impedance element may comprise a capacitor or a resistor. Typically, the sensing circuitry comprises DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit. The bias signal generator may comprises a voltage divider circuit connected between a reference voltage and a supply voltage of the controller. 
     The device is preferably a battery operated device and comprises one or more batteries for powering the device and further comprising an active fuse circuit for protecting the one or more batteries. The active fuse circuit may comprise a switch electrically coupled between a terminal of the one or more batteries and the drive circuit and control circuitry configured to open the switch to isolate the supply terminal from the drive circuit. 
     The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; a drive circuit for generating a cyclically varying drive signal for driving energy into the vessel/tissue; sensing circuitry for sensing a drive signal generated by the drive circuit; and a controller responsive to the sensing circuitry and operable to control the drive circuit to control the energy delivered to the vessel/tissue; wherein the drive circuit comprises an impedance element that is coupled to a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure. 
     The impedance element may be a capacitor or a resistor. The sensing circuitry may also comprise DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit. The bias signal generator may comprise a divider circuit connected between a reference voltage and a supply voltage of the controller. The divider circuit of the bias signal generator may be connected to the supply voltage of the controller via a switch and wherein the controller is configured to open the switch when the controller does not require signals from the sensing circuitry. 
     The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; one or more batteries for providing a DC voltage supply; a signal generator coupled to the one or more batteries for generating a cyclically varying drive signal from the DC voltage supply for driving energy into the vessel/tissue; a controller operable to control the signal generator to control the energy delivered to the vessel/tissue; and an active fuse circuit coupled between the one or more batteries and the signal generator for protecting the one or more batteries. 
     The active fuse circuit may comprise a switch that is electrically coupled between a terminal of the one or more batteries and the signal generator; and control circuitry configured to switch the switch. The switch may be arranged to disconnect the signal generator from the one or more batteries or may connect a large impedance between the signal generator and the one or more batteries. 
     The control circuitry of the active fuse may comprise circuitry for sensing a measure of the current being drawn from the one or more batteries and is configured to switch the switch in the event that the current measure exceeds a threshold. The control circuitry of the active fuse may comprise a comparator for comparing the current measure with the threshold and wherein an output of the comparator controls the opening and closing of the switch. 
     The present invention also provides a method of operating a medical device comprising generating a periodic drive signal and applying the drive signal to an ultrasonic transducer or to an end effector of the medical device and controlling the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector. 
     The present invention also provides a method of cauterising or cutting a vessel or tissue, the method comprising: gripping the vessel or tissue with an end effector of a medical device; using a drive circuit to apply a periodic drive signal either to an ultrasonic transducer or to the end effector; and controlling the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector. The method may use the above described medical device. 
     The present invention also provides electronic apparatus for use in a medical device having an ultrasonic transducer and an end effector, the electronic apparatus comprising: a drive circuit for generating a periodic drive signal; and a controller operable to control the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector; wherein the drive circuit comprise a first resonant circuit having a first resonant frequency and a second resonant circuit having a second resonant frequency that is different to the first resonant frequency, and wherein the controller is operable to control the drive circuit so that the drive circuit generates a drive signal having a frequency corresponding to the first resonant frequency when the drive signal is to be applied to the ultrasonic transducer and so that the drive circuit generates a drive signal having a frequency corresponding to the second resonant frequency when the drive signal is to be applied to the end effector. 
     The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; a drive circuit coupled to the end effector and operable to generate a drive signal and to provide the drive signal to the end effector; a controller operable to generate and output control signals to the drive circuit to control the drive signal generated by the drive circuit; wherein the drive circuit and a load formed by the vessel/tissue gripped by the end effector define a resonant circuit whose resonant frequency varies as the impedance of the load formed by the vessel/tissue gripped by the end effector changes; wherein the controller is arranged to generate control signals which cause the drive circuit to generate a drive signal having a frequency that tracks said resonant frequency as it changes; and wherein said controller is further arranged to reduce one or more of the power, current or voltage delivered to the load formed by the vessel/tissue gripped by the end effector. 
     Sensor circuitry may be provided for sensing signals applied to the load formed by the vessel/tissue gripped by the end effector and measurement circuitry for processing the signals from the sensor circuitry to determine a measure of the impedance of the load formed by the vessel/tissue gripped by the end effector. In this case, the controller can generate said control signals in dependence upon said measure of the impedance of the load formed by the vessel/tissue gripped by the end effector. 
     In one embodiment, the controller generates control signals having sequences of pulses and the controller skips one or more pulses from the control signals in order to reduce one or more of the power, current or voltage delivered to the load formed by the vessel/tissue gripped by the end effector. 
     Typically, in this case, the controller comprises a pulse signal generator that generates pulses at a desired frequency that depends on said resonant frequency and the controller skips pulses generated by said pulse signal generator by suppressing pulses generated by the pulse signal generator. 
    
    
     
       These and various other features and aspects of the invention will become apparent from the following detailed description of embodiments which are described with reference to the accompanying Figures in which: 
         FIG. 1  illustrates a hand-held cauterization device that has batteries and drive and control circuitry mounted in a handle portion of the device; 
         FIG. 2  is a part block diagram illustrating the main components of the cauterization device used in one embodiment of the invention; 
         FIG. 3  is a circuit diagram illustrating the main electrical components of the cauterization device shown in  FIG. 2 ; 
         FIG. 4  schematically illustrates the way in which the ultrasonic transducer is coupled to a waveguide for delivering the generated ultrasonic energy to the forceps and illustrating the way in which the circuitry shown in  FIG. 3  can deliver electrical energy to the forceps; 
         FIG. 5  is a block diagram that schematically illustrates processing modules that form part of the microprocessor shown in  FIG. 2 ; 
         FIG. 6  illustrates the form of control signals generated by the microprocessor to control the drive circuit whilst minimising 3 rd  harmonic content; 
         FIG. 7  is a contour plot illustrating the delivered power versus the load resistance and the drive frequency; 
         FIG. 8 a    is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor; 
         FIG. 8 b    is a circuit diagram illustrating another way in which a measure of the load current can be determined and supplied to the microprocessor; 
         FIG. 8 c    is a circuit diagram illustrating one way in which a measure of the load voltage can be determined and supplied to the microprocessor; 
         FIG. 9 a    is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor without using an op-amp circuit; 
         FIG. 9 b    is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor and illustrating one way in which a measure of the load voltage can be determined and supplied to the microprocessor without using op-amp circuits; and 
         FIG. 10  is a circuit diagram illustrating an active fuse circuit used to protect the batteries shown in  FIG. 2  from excessive current demand. 
     
    
    
     MEDICAL DEVICE 
     Many surgical procedures require cutting or ligating blood vessels or other vascular tissue. With minimally invasive surgery, surgeons perform surgical operations through a small incision in the patient&#39;s body. As a result of the limited space, surgeons often have difficulty controlling bleeding by clamping and/or tying-off transected blood vessels. By utilizing ultrasonic-surgical forceps or electro-surgical forceps, a surgeon can cauterize, coagulate/desiccate, and/or simply reduce bleeding by controlling the ultrasonic energy applied to the tissue/vessel by an ultrasonic transducer or by controlling the RF energy applied to the tissue/vessel via the forceps. 
       FIG. 1  illustrates the form of an ultrasonic/RF-surgical medical device  1  that is designed for minimally invasive medical procedures, according to one embodiment of the present invention. As shown, the device  1  is a self contained device, having an elongate shaft  3  that has a handle  5  connected to the proximal end of the shaft  3  and an end effector  7  connected to the distal end of the shaft  3 . In this embodiment, the end effector  7  comprises medical forceps  9  that are controlled by the user manipulating control levers  11  and  13  of the handle  5 . 
     During a surgical procedure, the shaft  3  is inserted through a trocar to gain access to the patient&#39;s interior and the operating site. The surgeon will manipulate the forceps  9  using the handle  5  and the control levers  11  and  13  until the forceps  9  are located around the vessel to be cut or cauterised. Electrical energy is then applied, in a controlled manner, either to the tissue directly via the forceps  9  (as RF energy) or to an ultrasonic transducer  8  that is mounted within the handle  5  and coupled to the forceps  9  via a waveguide (not shown) within the shaft  3 , in order to perform the desired cutting/cauterisation using ultrasonic energy. As shown in  FIG. 1 , in this embodiment, the handle  5  also houses batteries  15  and control electronics  17  for generating and controlling the electrical energy required to perform the cauterisation. In this way, the device  1  is self contained in the sense that it does not need a separate control box and supply wire to provide the electrical energy to the forceps  9 . However, such a separate control box may be provided if desired. 
     System Circuitry 
       FIG. 2  is a schematic block diagram illustrating the main electrical circuitry of the cauterization/cutting device  1  used in this embodiment to generate and control the electrical energy supplied to the ultrasonic transducer or to the forceps  9 . As will be explained in more detail below, in this embodiment, the circuitry is designed to control the period of an electrical drive waveform that is generated in order to control the amount of power delivered to the tissue/vessel being cauterized. 
     As shown in  FIG. 2 , the cauterization/cutting device  1  comprises a user interface  21 —via which the user is provided with information (such as an indication that energy is being applied to the gripped tissue/vessel by electrical energy or ultrasonic energy) and through which the user controls the operation of the cauterization/cutting device  1 , including selection of ultrasonic operation or RF operation. As shown, the user interface  21  is coupled to a microprocessor  23  that controls the cutting/cauterisation procedure by generating control signals that it outputs to gate drive circuitry  25 . In response to the control signals from the microprocessor  23 , the gate drive circuitry  25  generates gate control signals that cause a bridge signal generator  27  to generate a desired drive waveform that is applied either to the ultrasonic transducer  8  or to the forceps  9  via a drive circuit  29 . Voltage sensing circuitry  31  and current sensing circuitry  33  generate measures of the current and voltage applied to the ultrasonic transducer  8  or to the forceps  9 , which they feed back to the microprocessor  23  for control purposes.  FIG. 2  also shows the batteries  15  that provide the power for powering the electrical circuitry shown in  FIG. 2 . In this embodiment, the batteries  15  are arranged to supply 0V and 14V rails. 
       FIG. 3  illustrates in more detail the components of the gate drive circuitry  25 , the bridge signal generator  27  and the drive circuit  29 .  FIG. 3  also shows an electrical equivalent circuit  30  of the piezo-electric ultrasonic transducer  8  and the load (R load ) formed by the tissue/vessel to be treated. As shown in  FIG. 3 , the gate drive circuitry  25  includes two FET gate drives  37 —FET gate drive  37 - 1  and FET gate drive  37 - 2 . A first set of control signals (CTRL 1 ) from the microprocessor  23  is supplied to FET gate drive  37 - 1  and a second set of control signals (CTRL 2 ) from the microprocessor  23  is supplied to FET gate drive  37 - 2 . FET gate drive  37 - 1  uses the first set of control signals (CTRL 1 ) to generate two drive signals—one for driving each of the two FETs  41 - 1  and  41 - 2  of the bridge signal generator  27 . The FET gate drive  37 - 1  generates drive signals that causes the upper FET ( 41 - 1 ) to be on when the lower FET ( 41 - 2 ) is off and vice versa. This causes the node A to be alternately connected to the 14V rail (when FET  41 - 1  is switched on) and the 0V rail (when the FET  41 - 2  is switched on). Similarly, FET gate drive  37 - 2  uses the second set of control signals (CTRL 2 ) to generate two drive signals—one for driving each of the two FETs  41 - 3  and  41 - 4  of the bridge signal generator  27 . The FET gate drive  37 - 2  generates drive signals that causes the upper FET ( 41 - 3 ) to be on when the lower FET ( 41 - 4 ) is off and vice versa. This causes the node B to be alternately connected to the 14V rail (when FET  41 - 3  is switched on) and the 0V rail (when the FET  41 - 4  is switched on). Thus the two sets of control signals (CTRL 1  and CTRL 2 ) output by the microprocessor  23  control the digital waveform that is generated and applied between nodes A and B. Each set of control signals (CTRL 1  and CTRL 2 ) comprises of a pair of signal lines, one to indicate when the high side FET is on and the other to indicate when the low side FET is on. Thus the microprocessor  23 , either through software or through a dedicated hardware function can ensure that the undesirable condition when both high and low side FETs are simultaneously turned on does not occur. In practice this requires leaving a dead time when both high and low side FETs are turned off to ensure that, even when allowing for variable switching delays, there is no possibility that both FETs can be simultaneously on. In the present embodiment a dead time of about 100 ns was used. 
     As shown in  FIG. 3 , the nodes A and B are connected to the drive circuit  29 , thus the digital voltage generated by the bridge signal generator  27  is applied to the drive circuit  29 . This applied voltage will cause current to flow in the drive circuit  29 . As shown in  FIG. 3 , the drive circuit  29  includes two transformer circuits  42 - 1  and  42 - 2 . The first transformer circuit  42 - 1  is designed for efficient driving of the ultrasonic transducer  8  and includes a capacitor-inductor-inductor resonant circuit  43 - 1  formed by capacitor C US   s    45 , inductor L US   s    47  and inductor L US   m    49 . When driving the ultrasonic transducer  8 , the microprocessor  23  is arranged to generate control signals for the gate drive circuitry  25  so that the fundamental frequency (f d ) of the digital voltage applied across nodes A and B is around the resonant frequency of the resonant circuit  43 - 1 , which in this embodiment is about 50 kHz. As a result of the resonant characteristic of the resonant circuit  43 - 1 , the digital voltage applied across nodes A and B will cause a substantially sinusoidal current at the fundamental frequency (f d ) to flow within the resonant circuit  43 - 1 . This is because higher harmonic content of the drive voltage will be attenuated by the resonant circuit  43 - 1  and the impedance of L t  and C t1  referred to the transformer primary. 
     As illustrated in  FIG. 3 , the inductor L US   m    49  forms the primary of the transformer circuit  42 - 1 , the secondary of which is formed by inductor L US   sec    53 . The transformer up-converts the drive voltage (V US   d ) across the inductor L m    49  to a load voltage (V L ; typically about 120 volts) that is applied to the ultrasonic transducer  8 . The electrical characteristics of the ultrasonic transducer  8  change with the impedance of the forceps&#39; jaws and any tissue or vessel gripped by the forceps  9 ; and  FIG. 3  models the ultrasonic transducer  8  and the impedance of the forceps&#39; jaws and any tissue or vessel gripped by the forceps  9  by the inductor L t    57 , the parallel capacitors C t1    59  and C t2    61  and the resistance R load . 
     The inductor L US   s  and capacitor C US   s  of the drive circuit  29  are designed to have a matching LC product to that of inductor L t  and capacitor C t1  of the ultrasonic transducer  8 . Matching the LC product of a series LC network ensures that the resonant frequency of the network is maintained. Similarly, the magnetic reactance of the inductor L US   m  is chosen so that at resonance it matches with the capacitive reactance of the capacitor C t2  of the ultrasonic transducer  8 . For example, if the transducer  8  is defined such that capacitor C t2  has a capacitance of about 3.3 nF, then the inductor L US   m  should have an inductance of about 3 mH (at a resonant frequency of about 50 kHz). Designing the drive circuit  29  in this way provides for the optimum drive efficiency in terms of energy delivery to the tissue/vessel gripped by the forceps  9 . The efficiency improvement is realised because the current flowing in C US   s  and consequently the FET bridge ( 27 ) is reduced, because the transformer magnetising current cancels out the current flowing in C t2  In addition, because of this current cancellation, the current flowing in C US   s  is proportional to the current flowing in Rload, which allows the load current to be determined by measuring the current flowing in C US   s . 
     The second transformer circuit  42 - 2  is designed for efficient driving of electrical RF energy directly to the tissue/vessel via the forceps  9  and includes a capacitor-inductor-inductor resonant circuit  43 - 2  formed by capacitor C F   s    46 , inductor L F   s    48  and inductor L F   m    50 . When driving the forceps  9  directly with electrical energy, the microprocessor  23  is arranged to generate control signals for the gate drive circuitry  25  so that the fundamental frequency (f d ) of the digital voltage applied across nodes A and B is around the resonant frequency of the resonant circuit  43 - 2 , which in this embodiment is about 500 kHz. As a result of the resonant characteristic of the resonant circuit  43 - 2 , the digital voltage applied across nodes A and B will cause a substantially sinusoidal current at the fundamental frequency (f d ) to flow within the resonant circuit  43 - 2 . This is because higher harmonic content of the drive voltage will be attenuated by the resonant circuit  43 - 2 . 
     As illustrated in  FIG. 3 , the inductor L F   m    50  forms the primary of the transformer circuit  42 - 1 , the secondary of which is formed by inductor L US   sec    54 . The transformer up-converts the drive voltage (V F   d ) across inductor L F   m    50  to the load voltage (V F   L ; typically about 120 volts) that is applied to the forceps  9 . The tissue or vessel gripped by the jaws of the forceps  9  is represented as the resistive load R load  in the box labelled  9  in  FIG. 3 . In practice, this will be the same resistive load that is illustrated in the electrical equivalent circuit  30  of the ultrasonic transducer  8 . 
       FIG. 4  is a schematic diagram illustrating the way in which the ultrasonic transducer  8  couples to the tissue/vessel to be cauterized and the way in which the circuit components illustrated in  FIG. 3  connect to the ultrasonic transducer  8  and to the forceps  9 . In particular,  FIG. 4  shows the shaft  3 , the forceps  9  and the ultrasonic transducer  8 .  FIG. 4  also shows the waveguide  72  along which the ultrasonic signal that is generated by the ultrasonic transducer  8  is guided. The waveguide  72  is connected to the node “BB” show in  FIG. 3 , whilst the input supply to the ultrasonic transducer  8  is connected to node “AA” shown in  FIG. 3 . The output node “CC” of the second transformer circuit  42 - 2  is connected to a conductive inner wall of the sheath  3 , which is electrically connected to the upper jaw  74  of the forceps  9 . The return path is through the tissue/vessel to be cauterized and the lower jaw  76 , which is electrically connected to the node “BB”. 
     When the drive signal has a drive frequency of about 50 kHz, very little current will flow within the second transformer circuit  42 - 2  because the drive frequency is far away from the resonant frequency of the resonant circuit  43 - 2  such that the input impedance of the second transformer circuit  42 - 2  will be very high for this drive signal. Therefore, the power will be delivered almost entirely via the first transformer circuit  42 - 1 . Similarly, when the drive signal has a drive frequency of about 500 kHz, very little current will flow within the first transformer circuit  42 - 1  because the drive frequency is far away from the resonant frequency of the resonant circuit  43 - 1  such that the input impedance of the first transformer circuit  42 - 1  will be very high for this drive signal. Therefore, the power will be delivered almost entirely via the second transformer circuit  42 - 2 . In this way, the two transformer circuits  42 - 1  and  42 - 2  can be driven by a common bridge signal generator  27 ; although it is also feasible to drive each transformer circuit with separate bridge signal generators. 
     It is not always desired to apply full power to the tissue/vessel to be treated. Therefore, in this embodiment in the ultrasonic mode of operation, the amount of ultrasonic energy supplied to the vessel/tissue is controlled by varying the period of the digital waveform applied across nodes A and B so that the drive frequency (f d ) moves away from the resonant frequency of the ultrasonic transducer  8 . This works because the ultrasonic transducer  8  acts as a frequency dependent (lossless) attenuator. The closer the drive signal is to the resonant frequency of the ultrasonic transducer  8 , the more ultrasonic energy the ultrasonic transducer  8  will generate. Similarly, as the frequency of the drive signal is moved away from the resonant frequency of the ultrasonic transducer  8 , less and less ultrasonic energy is generated by the ultrasonic transducer  8 . In addition or instead, the duration of the pulses of the drive signals may be varied to control the amount of ultrasonic energy delivered to the tissue/vessel. 
     Similarly, in the electrical mode of operation, the amount of electrical power supplied to the forceps  9  is controlled by varying the period of the digital waveform applied across nodes A and B so that the drive frequency (f d ) moves away from the resonant frequency of the resonant circuit  43 - 2 . This works because the resonant circuit  43 - 1  acts as a frequency dependent (lossless) attenuator. The closer the drive signal is to the resonant frequency of the resonant circuit  43 - 1 , the less the drive signal is attenuated. Conversely, as the frequency of the drive signal is moved away from the resonant frequency of the circuit  43 - 1 , the more the drive signal is attenuated and so the electrical energy supplied to the tissue/vessel reduces. The drive frequency needs to move away from the resonant frequency by about 50% of the resonant frequency to achieve the desired range of power variation. An alternative approach to controlling the power, current or voltage applied during the electrical mode of operation is to continuously tune the frequency of the excitation signal to keep it matched with the resonant frequency of the drive circuit (as it changes with the changing load impedance) and thereby maintain efficient operation and to skip some of the pulses of the drive control signals until the average power, current and/or voltage is below the relevant limit. 
     A further alternative, which is most effective when using the pulse skipping technique, is to remove the inductor  48  shown in  FIG. 3  and therefore the drive circuit for the electrical mode of operation becomes a substantially parallel LC resonant circuit (it is not a pure parallel LC resonant circuit because the transformer leakage inductance appears in series with inductor  48  and cannot be entirely removed). The advantage of removing the inductor  48  is that the overall efficiency can be increased, because there are no longer any losses in the inductor. A further advantage is that the physical size of the circuit can be reduced, because often the inductor is a physically large component relative to the FETs, microprocessor, capacitors and other system components. 
     The microprocessor  23  controls the power delivery based on a desired power to be delivered to the circuitry  30  (which models the ultrasonic transducer  8  and the tissue/vessel gripped by the forceps  9 ) or to the forceps  9  and based on measurements of the load voltage (V L ) and of the load current (i L ) obtained from the voltage sensing circuitry  31  and the current sensing circuitry  33 . The microprocessor  23  also selects the frequency of the drive signal (around 50 kHz or around 500 kHz) based on a user input received via the user interface  21  that selects either electrical operation or ultrasonic operation. 
     Microprocessor 
       FIG. 5  is a block diagram illustrating the main components of the microprocessor  23  that is used in this embodiment. As shown, the microprocessor  23  includes synchronous I,Q sampling circuitry  81  that receives the sensed voltage and current signals from the sensing circuitry  31  and  33  and obtains corresponding samples which are passed to a measured voltage and current processing module  83 . The measured voltage and current processing module  83  uses the received samples to calculate the impedance of, and the RMS voltage applied to and the RMS current flowing through, the ultrasonic transducer  8  and/or directly to the tissue/vessel gripped by the forceps  9 ; and from them the power that is presently being supplied to the circuitry  30  or directly to the tissue/vessel gripped by the forceps  9 . The determined values are then passed to a power controller  85  for further processing. The measured voltage and current processing module  83  can also process the received I and Q samples to calculate the phase difference between the load voltage (V L ) and the load current (i L ). During the ultrasonic mode of operation, at resonance, this phase difference should be around zero and so this phase measure can be used as a feedback parameter for the power controller  85 . 
     The power controller  85  uses the received impedance value and the delivered power value to determine, in accordance with a predefined algorithm and a power set point value and a mode indication signal (received from a medical device control module  89  and indicating ultrasonic operation or electrical operation), a desired period/frequency (Δt new ) of the control signals (CTRL 1  and CTRL 2 ) that are used to control the gate drive circuit  25 . This desired period/frequency is passed from the power controller  85  to the control signal generator  95 , which changes the control signals CTRL 1  and CTRL 2  in order to change the waveform period to match the desired period. The CTRL control signals may comprise square wave signals having the desired period or they may comprise periodic pulses with the period corresponding to the desired period (Δt new ) and with the relative timing of the pulses of the control signals being set to minimise harmonic content of the waveform that is generated by the bridge signal generator  27  (such as to minimise the 3 rd  order harmonic). In this embodiment, the control signals CTRL 1  are output to the FET gate drive  37 - 1  (shown in  FIG. 2 ), which amplifies the control signals and then applies them to the FETs  41 - 1  and  41 - 2 ; and the control signals CTRL 2  are output to the FET gate drive  37 - 2  (shown in  FIG. 2 ), which amplifies the control signals and then applies them to the FETs  41 - 3  and  41 - 4 , to thereby generate the desired waveform with the new period (Δt new ). 
     In order to drive the circuitry with the optimal RMS waveform, the MOSFETs  41  are driven as complementary, opposing pairs. Although the maximum output voltage is achieved when the MOSFET pairs are driven at a phase shift of 180 degrees, the resulting harmonic content of such a drive waveform, particularly the 3rd harmonic (which is poorly excluded by the output filter) is quite high. The inventors have found that the optimal phase shift between the control signals applied to the two pairs of MOSFETs  41 , for 3rd harmonic reduction, is around 120°. This is illustrated in  FIG. 6  which shows in the upper plot the output from the first MOSFET pair  41 - 1  and  41 - 2 ; in the middle plot the output from the second MOSFET pair  41 - 3  and  41 - 4  (shifted by 120° relative to the upper plot); and in the lower plot the resulting (normalised) output voltage applied across inputs A and B. The shape of this normalised output voltage has very low 3 rd  order harmonic content. 
     I &amp; Q Signal Sampling 
     Both the load voltage and the load current will be substantially sinusoidal waveforms, although they may be out of phase, depending on the impedance of the load represented by the transducer  8  and/or the vessel/tissue gripped by the forceps  9 . The load current and the load voltage will be at the same drive frequency (f d ) corresponding to the presently defined waveform period (Δt new ). Normally, when sampling a signal, the sampling circuitry operates asynchronously with respect to the frequency of the signal that is being sampled. However, as the microprocessor  23  knows the frequency and phase of the switching signals, the synchronous sampling circuit  81  can sample the measured voltage/current signal at predefined points in time during the drive period. In this embodiment, during the ultrasonic mode of operation, the synchronous sampling circuit  81  oversamples the measured signal eight times per period to obtain four I samples and four Q samples. Oversampling allows for a reduction of errors caused by harmonic distortion and therefore allows for the more accurate determination of the measured current and voltage values. However, oversampling is not essential and indeed under sampling (less than two samples per period) is performed when the device is operating in the electrical mode of operation and is possible due to the synchronous nature of the sampling operation. The timing that the synchronous sampling circuit  81  makes these samples is controlled, in this embodiment, by the control signals CTRL 1  and CTRL 2 . Thus when the period of these control signals is changed, the period of the sampling control signals CTRL 1  and CTRL 2  also changes (whilst their relative phases stay the same). In this way, the sampling circuitry  81  continuously changes the timing at which it samples the sensed voltage and current signals as the period of the drive waveform is changed so that the samples are always taken at the same time points within the period of the drive waveform. Therefore, the sampling circuit  81  is performing a “synchronous” sampling operation instead of a more conventional sampling operation that just samples the input signal at a fixed sampling rate defined by a fixed sampling clock. Of course, such a conventional sampling operation could be used instead. 
     Measurements 
     The samples obtained by the synchronous sampling circuitry  51  are passed to the measured voltage and current processing module  83  which can determine the magnitude and phase of the measured signal from just one “I” sample and one “Q” sample of the load current and load voltage. However, in this embodiment, to achieve some averaging, the processing module  83  averages consecutive “I” samples to provide an average “I” value and consecutive “Q” samples to provide an average “Q” value; and then uses the average I and Q values to determine the magnitude and phase of the measured signal. Of course, it should be recognised that some pre-processing of the data may be required to convert the actual measured I and Q samples into I and Q samples of the load voltage or the load current, for example, scaling, integration or differentiation of the sample values may be performed to convert the sampled values into true samples of the load voltage (V L ) and the load current (i L ). Where integration or differentiation is required, this can be achieved simply by swapping the order of the I and Q samples—as integrating/differentiating a sinusoidal signal simply involves a scaling and a 90 degree phase shift. 
     The RMS load voltage, the RMS load current and the delivered power, P delivered , can then be determined from: 
     
       
         
           
             
               V 
               RMS 
             
             = 
             
               
                 1 
                 
                   2 
                 
               
                
               
                 
                   ( 
                   
                     
                       V 
                       I 
                       2 
                     
                     + 
                     
                       V 
                       Q 
                       2 
                     
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             
               I 
               RMS 
             
             = 
             
               
                 1 
                 
                   2 
                 
               
                
               
                 
                   ( 
                   
                     
                       I 
                       I 
                       2 
                     
                     + 
                     
                       I 
                       Q 
                       2 
                     
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             Power 
             = 
             
               
                 V 
                 · 
                 
                   I 
                   * 
                 
               
               = 
               
                 
                   
                     1 
                     
                       2 
                     
                   
                    
                   
                     ( 
                     
                       
                         V 
                         I 
                       
                       + 
                       
                         j 
                          
                         
                           V 
                           Q 
                         
                       
                     
                     ) 
                   
                    
                   
                     ( 
                     
                       
                         I 
                         I 
                       
                       - 
                       
                         jI 
                         Q 
                       
                     
                     ) 
                   
                 
                 = 
                 
                   
                     P 
                     delivered 
                   
                   + 
                   
                     j 
                      
                     
                       P 
                       reactive 
                     
                   
                 
               
             
           
         
       
       
         
           
             
               P 
               delivered 
             
             = 
             
               
                 1 
                 
                   2 
                 
               
                
               
                 ( 
                 
                   
                     
                       V 
                       I 
                     
                      
                     
                       I 
                       I 
                     
                   
                   + 
                   
                     
                       V 
                       Q 
                     
                      
                     
                       I 
                       Q 
                     
                   
                 
                 ) 
               
             
           
         
       
       
         
           
             
               P 
               reactive 
             
             = 
             
               
                 1 
                 
                   2 
                 
               
                
               
                 ( 
                 
                   
                     
                       V 
                       Q 
                     
                      
                     
                       I 
                       I 
                     
                   
                   - 
                   
                     
                       V 
                       I 
                     
                      
                     
                       I 
                       Q 
                     
                   
                 
                 ) 
               
             
           
         
       
       
         
           
             
                
               Power 
                
             
             = 
             
               
                 
                   V 
                   RMS 
                 
                  
                 
                   I 
                   RMS 
                 
               
               = 
               
                  
                 
                   
                     P 
                     delivered 
                   
                   + 
                   
                     j 
                      
                     
                       P 
                       reactive 
                     
                   
                 
                  
               
             
           
         
       
     
     The impedance of the load represented by the ultrasonic transducer  8  and the vessel/tissue gripped by the forceps  9  (or just the impedance of the forceps  9  and the vessel/tissue gripped by the forceps  9  if the electrical energy is directly applied to the forceps  9 ) can be determined from: 
     
       
         
           
             
               Z 
               Load 
             
             = 
             
               
                 
                   ( 
                   
                     
                       V 
                       I 
                     
                     + 
                     
                       j 
                        
                       
                         V 
                         Q 
                       
                     
                   
                   ) 
                 
                 
                   ( 
                   
                     
                       I 
                       I 
                     
                     + 
                     
                       jI 
                       Q 
                     
                   
                   ) 
                 
               
               = 
               
                 
                   
                     
                       ( 
                       
                         
                           V 
                           I 
                         
                         + 
                         
                           jV 
                           Q 
                         
                       
                       ) 
                     
                      
                     
                       ( 
                       
                         
                           I 
                           I 
                         
                         - 
                         
                           jI 
                           Q 
                         
                       
                       ) 
                     
                   
                   
                     
                       ( 
                       
                         
                           I 
                           I 
                         
                         + 
                         
                           jI 
                           Q 
                         
                       
                       ) 
                     
                      
                     
                       ( 
                       
                         
                           I 
                           I 
                         
                         - 
                         
                           jI 
                           Q 
                         
                       
                       ) 
                     
                   
                 
                 = 
                 
                   
                     
                       ( 
                       
                         
                           
                             V 
                             I 
                           
                            
                           
                             I 
                             I 
                           
                         
                         + 
                         
                           
                             V 
                             Q 
                           
                            
                           
                             I 
                             Q 
                           
                         
                         + 
                         
                           
                             jV 
                             Q 
                           
                            
                           
                             I 
                             I 
                           
                         
                         - 
                         
                           
                             jjV 
                             I 
                           
                            
                           
                             I 
                             Q 
                           
                         
                       
                       ) 
                     
                     
                       
                         2 
                       
                        
                       
                         I 
                         RMS 
                         2 
                       
                     
                   
                   = 
                   
                     
                       R 
                       Load 
                     
                     + 
                     
                       jX 
                       Load 
                     
                   
                 
               
             
           
         
       
     
     An alternative way of computing R Load  and X Load  is as follows: 
     
       
         
           
             
               R 
               Load 
             
             = 
             
               
                 
                   
                     P 
                     delivered 
                   
                   
                     
                       2 
                     
                      
                     
                       I 
                       
                         R 
                          
                         M 
                          
                         S 
                       
                       2 
                     
                   
                 
                  
                 
                     
                 
                  
                 
                   X 
                   Load 
                 
               
               = 
               
                 
                   P 
                   reactive 
                 
                 
                   
                     2 
                   
                    
                   
                     I 
                     RMS 
                     2 
                   
                 
               
             
           
         
       
     
     and the phase difference between the load voltage and the load current can be determined from: 
       Phase measured   =a  tan 2( P   reactive   ,P   delivered ) 
     A computationally efficient, approximation to the a tan 2 function can be made using look up tables and interpolation in fixed point arithmetic, or using a ‘CORDIC’ like algorithm, 
     Limits 
     As with any system, there are certain limits that can be placed on the power, current and voltage that can be delivered either to the ultrasonic transducer  8  or to the forceps  9 . The limits used in this embodiment and how they are controlled will now be described. 
     In this embodiment, the drive circuitry  29  is designed to deliver ultrasonic energy into tissue or to deliver electrical energy into tissue with the following requirements:
     1) Supplied with a nominally 14V DC supply   2) Substantially sinusoidal output waveform at approximately 50 kHz in the case of ultrasonic operation   3) Substantially sinusoidal output waveform at approximately 500 kHz in the case of RF electrical operation   4) Power limited output of 90 W in the case of ultrasonic operation   5) Power limited output of 100 W in the case of electrical operation   6) Current limited to 1.4A rms  and voltage limited to 130V rms  in the case of ultrasonic operation   7) Current limited to 1.4A rms  and voltage limited to 100V rms  in the case of electrical operation   8) In the case of ultrasonic operation, the measured phase is greater than a system defined phase limit   

     The power controller  85  maintains data defining these limits and uses them to control the decision about whether to increase or decrease the waveform period or whether to skip pulses of the control signals given the latest measured power, load impedance and/or measured phase. In this embodiment, when operating in the ultrasonic mode of operation, the phase limit that is used depends on the measured load impedance. In particular, the power controller  85  maintains a look up table (not shown) relating load impedance to the phase limit; and the values in this table limit the phase so that when the measured load impedance is low (indicating that the jaws of the forceps  9  are open and not gripping tissue or a vessel), the delivered power is reduced (preferably to zero). 
     As discussed above, one of the ways to control the operation of the device (when operating in the electrical mode of operation) is to maximise the drive efficiency. When controlling the device in this way, the power controller  85  tracks a maximum power delivery condition as the load changes. The way that this can be done will now be described. 
     Maximum Power Delivery Tracking Condition 
     The complex impedance of the circuitry shown in  FIG. 3  (when operating in the electrical mode of operation and with inductor  48  removed) can be approximated by the following equation: 
     
       
         
           
             Z 
             = 
             
               
                 j 
                  
                 
                     
                 
                  
                 2 
                  
                 
                     
                 
                  
                 π 
                  
                 
                     
                 
                  
                 
                   fL 
                   S 
                   F 
                 
               
               + 
               
                 1 
                 
                   J 
                    
                   
                       
                   
                    
                   2 
                    
                   
                       
                   
                    
                   π 
                    
                   
                       
                   
                    
                   
                     fC 
                     S 
                     F 
                   
                 
               
               + 
               
                 
                   j 
                    
                   
                       
                   
                    
                   2 
                    
                   
                       
                   
                    
                   π 
                    
                   
                       
                   
                    
                   
                     fL 
                     M 
                     F 
                   
                    
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                   
                 
                 
                   
                     j 
                      
                     
                         
                     
                      
                     2 
                      
                     
                         
                     
                      
                     π 
                      
                     
                         
                     
                      
                     
                       fL 
                       M 
                       F 
                     
                   
                   + 
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                   
                 
               
               + 
               
                 R 
                 s 
               
             
           
         
       
     
     Where: 
     R load_ref  is the load resistance referred to the primary (by the square of the turns ratio); and R s  represents the equivalent series resistance of the inductor, transformer capacitor and switching devices. This complex impedance may be rewritten as: 
     
       
         
           
             Z 
             = 
             
               
                 j 
                  
                 
                     
                 
                  
                 2 
                  
                 
                     
                 
                  
                 π 
                  
                 
                     
                 
                  
                 
                   fL 
                   S 
                   F 
                 
               
               + 
               
                 1 
                 
                   J 
                    
                   
                       
                   
                    
                   2 
                    
                   
                       
                   
                    
                   π 
                    
                   
                       
                   
                    
                   
                     fC 
                     S 
                     F 
                   
                 
               
               + 
               
                 
                   
                     4 
                      
                     
                       π 
                       2 
                     
                      
                     
                       f 
                       2 
                     
                      
                     
                       L 
                       M 
                       
                         F 
                         2 
                       
                     
                      
                     
                         
                       
                         R 
                         
                           load 
                            
                           _ 
                            
                           ref 
                         
                       
                     
                   
                   
                     4 
                      
                     
                       π 
                       2 
                     
                      
                     
                       f 
                       2 
                     
                      
                     
                       L 
                       M 
                       
                         F 
                         2 
                       
                     
                      
                     
                         
                       
                         + 
                         
                           R 
                           
                             load 
                              
                             _ 
                              
                             ref 
                           
                           2 
                         
                       
                     
                   
                 
                  
                 
                   
                     j 
                      
                     
                         
                     
                      
                     2 
                      
                     
                         
                     
                      
                     π 
                      
                     
                         
                     
                      
                     
                       fL 
                       M 
                       F 
                     
                      
                     
                         
                       
                         R 
                         
                           load 
                            
                           _ 
                            
                           ref 
                         
                         2 
                       
                     
                   
                   
                     4 
                      
                     
                       π 
                       2 
                     
                      
                     
                       f 
                       2 
                     
                      
                     
                       L 
                       M 
                       
                         F 
                         2 
                       
                     
                      
                     
                         
                       
                         + 
                         
                           R 
                           
                             load 
                              
                             _ 
                              
                             ref 
                           
                           2 
                         
                       
                     
                   
                 
               
               + 
               
                 R 
                 s 
               
             
           
         
       
     
     Therefore, the real part of this complex impedance is: 
     
       
         
           
             
                
               
                 ( 
                 Z 
                 ) 
               
             
             = 
             
               
                 
                   4 
                    
                   
                     π 
                     2 
                   
                    
                   
                     f 
                     2 
                   
                    
                   
                     L 
                     M 
                     
                       F 
                        
                       
                           
                       
                        
                       2 
                     
                   
                    
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                   
                 
                 
                   
                     4 
                      
                     
                       π 
                       2 
                     
                      
                     
                       f 
                       2 
                     
                      
                     
                       L 
                       M 
                       
                         F 
                          
                         
                             
                         
                          
                         2 
                       
                     
                   
                   + 
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                     2 
                   
                 
               
               + 
               
                 R 
                 s 
               
             
           
         
       
     
     And the imaginary part of this complex impedance is: 
     
       
         
           
             
                
               
                 ( 
                 Z 
                 ) 
               
             
             = 
             
               
                 2 
                  
                 π 
                  
                 
                     
                 
                  
                 
                   fL 
                   S 
                   F 
                 
               
               - 
               
                 1 
                 
                   2 
                    
                   π 
                    
                   
                       
                   
                    
                   
                     fC 
                     S 
                     F 
                   
                 
               
               + 
               
                 
                   2 
                    
                   π 
                    
                   
                       
                   
                    
                   
                     fL 
                     M 
                     F 
                   
                    
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                     2 
                   
                 
                 
                   
                     4 
                      
                     
                       π 
                       2 
                     
                      
                     
                       f 
                       2 
                     
                      
                     
                       L 
                       M 
                       
                         F 
                          
                         
                             
                         
                          
                         2 
                       
                     
                   
                   + 
                   
                     R 
                     
                       load 
                        
                       _ 
                        
                       ref 
                     
                     2 
                   
                 
               
             
           
         
       
     
     When the drive frequency (t) corresponds to the resonant frequency of this complex impedance, the imaginary part  (Z)=0. Therefore, the power controller  85  can vary the drive frequency (t) to keep the imaginary part  (Z) at or around zero using a phase locked loop. Indeed, it can be shown that when  (Z)=0 the maximum power (for a given supply voltage) is delivered to the load. 
       FIG. 7  is a contour plot showing the power contours than can be delivered to the load versus the drive frequency and the load resistance (Rload). As shown in  FIG. 7 , the power that can be delivered varies with the load resistance and the drive frequency.  FIG. 7  also shows the line  92  of maximum power delivery that can be achieved as the load resistance and drive frequency change. Therefore, the power controller  85  can use the measured value of Rload together with stored data defining the line  92  shown in  FIG. 7  (which may be a look-up-table) to determine the corresponding drive frequency to be used. In this way, the microprocessor  23  will track along the line  92  shown in  FIG. 7  as the load resistance changes during the cutting/cauterisation process. 
     One of the advantages of this approach is that it enables a useful operating condition at low values of Rload, in particular for values of Rload less than the critical value (i.e. when in which a maximum power will be delivered even if this is below the R load_ref &lt;2π∫L F   M ), desired power level. However, operating along the line  92  of maximum power delivery can result in some of the above system limits being breached unless further control action is taken. In the preferred embodiment, this further control action is to use pulse skipping techniques until the average power, current and/or voltage is below the relevant limit. For example, as can be seen from the measurements described above, the measured voltage and current processing module  83  can determine the delivered power, the RMS voltage and the RMS current. The power controller  85  can therefore use these values to skip one or more pulses of the CTRL control signals until the measured voltage and current values are below the relevant system limits and the delivered power is at or below the power set-point defined by the medical device control module  89 . 
     Pulses may be skipped, for example, by passing the pulses generated by the control signal generator  87  through a logic gate (not shown) and selectively suppressing pulses that are generated by the control signal generator  87  by controlling the logic level of another input to the logic gate. For example, the pulses of each control signal that is generated by the control signal generator  87  may be passed through an AND gate, with another input of the AND gate being generated by the power controller  85  and being a logic “1” when the pulses are to be output to the FET gate drives  37  as normal and being a logic “0” when the pulses are to be skipped or suppressed. Other pulse skipping techniques could of course be used. 
     Medical Device Control Module 
     As mentioned above, the medical device control module  89  controls the general operation of the cauterisation/cutting device  1 . It receives user inputs via the user input module  91 . These inputs may specify that the jaws of the forceps  9  are now gripping a vessel or tissue and that the user wishes to begin cutting/cauterisation and specify whether ultrasonic energy or electrical energy is to be applied to the vessel/tissue. In response, in this embodiment, the medical device control module  89  initiates a cutting/cauterisation control procedure. Initially, the medical device control module  89  sends an initiation signal to the power controller  85  and obtains the load impedance measurements determined by the measured voltage and current processing module  83 . The medical device control module  89  then checks the obtained load impedance to make sure that the load is not open circuit or short circuit. If it is not, then the medical device control module  89  starts to vary the power set point to perform the desired cutting/cauterisation and sets the initial period/frequency of the drive signal to be generated. As discussed above, for ultrasonic operation, the initial frequency of the drive signal will be set around 50 kHz and for RF electrical operation, the initial frequency will be set around 500 kHz. 
     Voltage/Current Sensing Circuitry 
     As shown in  FIG. 2 , voltage sensing circuitry  31  is provided to sense the load voltage applied to the load and current sensing circuitry  33  is provided to sense the current applied to the load. The sensed signals are supplied to the microprocessor  23  for use in controlling the operation of the medical device. There are various ways of sensing the load voltage and the load current and some of these will now be described. 
       FIG. 8 a    illustrates the primary side of the first transformer circuit  42 - 1  and one way in which the current sensing circuitry  33  obtains a measure of the load current. As shown, the current sensing circuitry  33  comprises an additional inductor turn(s)  67  which link the flux present in inductor  47  (or inductor  49 ) and that consequently outputs a voltage across inductor  67  that varies with the rate of change of load current. The voltage across the inductor  67  is a bipolar voltage whose amplitude is directly proportional to the rate of change of load current and the number of turns in  67 . This bipolar voltage is scaled and converted into a unipolar voltage suitable for input to the microprocessor  23  by the op-amp circuit  69 - 1  which outputs a measured voltage (V meas ). This measured voltage will also depend on the current flowing in the inductor  47  and so will also depend on the current flowing on the secondary side of the transformer circuit  42 - 1  and thus the current flowing through the load. As the ratio of the number of turns of inductor  47  to inductor  67  is known, the measured voltage and current processing module  83  can use V meas  to determine the voltage across inductor  47 . The voltage across inductor  47  is related to the current flowing through the inductor  47  by V=Ldi/dt. As the inductance of inductor  47  is known, the measured voltage and current processing module  83  can determine the current flowing in the primary side of the transformer circuit  42 - 1  by integrating the voltage across the inductor  47  and by scaling the result to account for the inductance of the inductor  47  (and the scaling of the op-amp-circuit  69 - 1 ). This current measure can then be converted into a suitable measure of the load current (i L ) by a further scaling to take into account the number of turns between inductor  49  and inductor  53 . Of course, the measured voltage and current processing module  83  does not need to integrate the voltage across the inductor  47 —as the measured signals are sinusoidal and so integration can be achieved by applying a suitable scaling factor and a 90 degree phase shift. Thus, the measured voltage and current processing module  83  can determine the load current by applying a suitable (pre-stored) scale factor to the measured voltage (V meas ) and by applying a suitable 90 degree phase shift (which can be achieved simply by swapping the order of the I and Q samples as discussed above). 
       FIG. 8 b    illustrates the primary side of the first transformer circuit  42 - 1  and another way in which the current sensing circuitry  33  can obtain a measure of the load current. As shown, in this case, the current sensing circuitry  33  measures the voltage across the capacitor  45 . The voltage across the capacitor  45  is a bipolar voltage. This bipolar voltage is scaled and converted into a unipolar voltage suitable for input to the microprocessor  23  by the op-amp circuit  69 - 2  which outputs a measured voltage (V meas ). This voltage is related to the current flowing in the primary side of the transformer circuit  42  by I=CdV meas /dt and thus the current flowing through the load. As the capacitance of the capacitor  45  is known, the measured voltage and current processing module  83  can determine the current flowing in the primary side of the transformer circuit  42 - 1  by differentiating the voltage across the capacitor  45  and by scaling the result to account for the capacitance of the capacitor  45  (and the scaling of the op-amp-circuit  69 - 2 ). This current measure can then be converted into a suitable measure of the load current (i L ) by a further scaling to take into account the number of turns between inductor  49  and inductor  53 . Of course, the measured voltage and current processing module  83  does not need to differentiate the voltage across the capacitor  45 —as the measured signals are sinusoidal and so differentiation can be achieved by applying a suitable scaling factor and a 90 degree phase shift. Thus, the measured voltage and current processing module  83  can determine the load current by applying a suitable (pre-stored) scale factor to the measured voltage (V meas ) and by applying a suitable 90 degree phase shift (which can be achieved simply by swapping the order of the I and Q samples as discussed above). 
       FIG. 8 c    schematically illustrates how a measure of the load voltage can be determined.  FIG. 8 c    shows the secondary side of the first transformer circuit  42  and illustrates the use of a voltage divider circuit (in this case formed by resistors R 1  and R 2 ), with the voltage across resistor R 2  being input to the op-amp circuit  69 - 3 . Therefore, by applying an appropriate scale on the measured voltage from op-amp  69 - 3 , the measured voltage and current processing module  83  can determine the load voltage. 
     The sensing circuits described above use op-amp circuits  69  to convert the bipolar drive signals into unipolar voltages that are suitable for input to the microprocessor  23 . The use of such op-amp circuits has a number of disadvantages, including that they are costly, they consume power and they requires space within the electronics. These are important factors when the device is designed to be battery powered and the electronics are housed within the handle  5  of the device.  FIG. 9  illustrates various sensing circuits that can be used without an op-amp. The circuit of  FIG. 9 a    is suitable when the AC drive signal is unipolar. This may be achieved by replacing the full bridge signal generator  27  with a half bridge signal generator. This would mean removing, for example, the FETs  41 - 3  and  41 - 4  and connecting node B to ground. In this case, the microprocessor  23  only needs to generate one control signal (CTRL 1 ) to control FETs  41 - 1  and  41 - 2 . The circuit of  FIG. 9 b    is suitable for both unipolar and bipolar drive signals. 
     In  FIG. 9 a   , the capacitor  45  has been moved to be between the inductor  49  that forms the primary of the transformer circuit  42 - 1  and ground (GND). The current sensing circuitry  33  is arranged to measure the voltage across the capacitor  45  via a potential divider formed by resistors R 1 //R 2  and R 3 . The resistor R 1  connects the output of a DC blocking capacitor C B  to a supply voltage rail of the microprocessor  23  (in this case at 3.3V) through the switch  121 ; and resistor R 2  connects the output of the DC blocking capacitor C B  to a reference potential (in this case ground). The resistors R 1  and R 2  therefore provide a divider circuit that applies a DC bias to the measured AC signal. The DC blocking capacitor prevents this DC bias from coupling to the drive circuit. Typically, the resistors R 1  and R 2  are equal so that the DC bias will be at 1.65V. Thus, the voltage output from the sensing circuit  33  will be an AC voltage having a mid-rail value of about 1.65 V and whose peak voltage will be a proportion of the voltage across the capacitor  45 . The potential divider formed by resistors R 1 //R 2  and R 3  is such that the peak to peak amplitude of the AC signal passed to the microprocessor  23  is less than the 3.3V input range of the microprocessor  23 . If the microprocessor  23  operates at a different voltage rail (for example 5V), then the values of the resistors R 1 , R 2  and R 3  can be adjusted accordingly. To minimise current drawn either from the 3.3V rail or from the transformer circuit  42 - 1 , the resistors R 1 , R 2  and R 3  can have relatively large values and typical values for these resistors are: R 1 =R 2 =200Ω and R 3 =1000Ω. The values of R 1  and R 2  should be chosen to meet the input impedance requirements of the Analogue to digital converter used to sample the signals. The switch  121  allows the microprocessor  23  to disconnect the sensing circuit  33  from the 3.3V rail—so that when sensing is not required, the circuit  33  does not consume any power. 
       FIG. 9 b    illustrates the way in which similar circuits can be provided on the secondary side of the transformer circuit  42 - 1 . In particular,  FIG. 9 b    shows the voltage sensing circuit  31  used to obtain a measure of the load voltage V L  via a voltage divider formed by capacitors C 1  and C 2 . If the capacitors C 1  and C 2  were replaced with resistors, then a blocking capacitor C B  should be provided before the divider circuit formed by resistors R 1  and R 2  in the manner shown in  FIG. 9 a   .  FIG. 9 b    also shows the current sensing circuit  33 , which senses the load current by sensing the voltage across capacitor C 3  via a potential divider formed by resistors R 4 //R 5  and R 6 . As shown, a DC blocking capacitor C B  is provided to enable a DC bias signal to be added to the output to the microprocessor via the divider circuit formed from resistors R 4  and R 5 . As before, the switches  121 - 1  and  121 - 2  allow the microprocessor to switch off the sensing circuits  31  and  33  when sensor signals are not required. As those skilled in the art will appreciate, the “op-ampless” sensing circuits  31  and  33  illustrated in  FIG. 9  are cheaper to manufacture, can be made to consume less power and take up less space on the circuit board than the op-amp circuits illustrated in  FIG. 8 . 
     The circuits illustrated in  FIGS. 8 and 9  were used to obtain measurements from the first transformer circuit  42 - 1  of the drive circuit  29 . As those skilled in the art will appreciate, the same or similar sensing circuits would be provided for sensing signals in the second transformer circuit  42 - 2 . Further, although the sensing circuits illustrated in  FIG. 9  sense the voltage across a capacitor, the circuits could also sense the voltage across another impedance element, such as across a resistor of the transformer circuit  42 . 
     Active Battery Protection 
     Typically, with battery operated devices such as the medical device  1  described above, a fuse is provided between the batteries and the electric circuitry, to protect the battery from damage caused by short circuits and the like. However, standard fuses have a resistance of about 10 mOhms. With such a standard fuse, when 10A is drawn from the batteries, approximately 1 W is dissipated through the fuse. An active fuse circuit  130  that is used in this embodiment will now be described with reference to  FIG. 10  that reduces the power dissipation associated with such standard fuses. 
       FIG. 10  shows the batteries  15 , which supply the 14V rail and the GND rail to the circuitry shown in  FIG. 3 . The active fuse circuit  130  comprises a differential amplifier  131  that measures the voltage drop across part of a PCB conductor trace  133  that is connected between the 14V rail and the positive terminal of the batteries (V bat+ ). The conductor trace  133  has a resistance of approximately 1-2 mOhms and so the voltage drop is proportional to the current that is being drawn from the batteries  15 . The measured voltage drop is low pass filtered by the low pass filter  135  to avoid transient spikes triggering the fuse circuit—so that it is just the voltage corresponding to the DC current drawn from the batteries  15  that will pass through the filter  135 . In this embodiment, the low pass filter  135  has a cut-off frequency of about 10 Hz. The output from the low pass filter  135  is then compared with a reference voltage (V ref ) using a latching comparator  137 . The reference voltage is set in advance to correspond to a desired limit on the current drawn from the batteries. In this embodiment, V ref  is set to correspond to a current limit of 15A. When the voltage drop across the trace  133  is less than the reference voltage the output of the comparator  137  remains at a high value—which maintains the FET switch  139  on and so current can be drawn from the batteries  15  by the bridge signal generator  27 . However, when the voltage drop across the trace  133  is greater than the reference voltage, then the output of the comparator  137  goes low and it is maintained low even if the current drawn from the batteries drops below the defined limit. When the comparator output is low the FET  139  is switched off, thereby disconnecting the batteries  15  from at least the bridge signal generator  27 . 
     In this embodiment, the FET  139  is an N-channel enhanced mode switch having an on resistance of just 2 mOhms. This means that when the switch  139  is switched on and 10A is being drawn from the batteries, just 0.2 W is dissipated through the switch  139 . 
     In this embodiment, when the comparator  137  is triggered and the switch  139  is switched off (open circuit), the batteries  15  have to be removed to reset the active fuse circuitry  130 —as the opening of the switch  139  disconnects all the electronics except the circuit components of the active fuse  130  from the batteries. Alternatively, if the microprocessor  23  (or some other control circuitry) is powered directly from the batteries, then the comparator  137  could be reset either in response to a user input (for example in response to the user pressing a reset button or the like) or in response to some other trigger event (such as after a predetermined time-out period). 
     Modifications and Alternatives 
     A medical cauterisation/cutting device has been described above. As those skilled in the art will appreciate, various modifications can be made and some of these will now be described. Other modifications will be apparent to those skilled in the art. 
     In the above embodiment, various operating frequencies, currents, voltages etc were described. As those skilled in the art will appreciate, the exact currents, voltages, frequencies, capacitor values, inductor values etc. can all be varied depending on the application and any values described above should not be considered as limiting in any way. However, in general terms, the circuit described above has been designed to provide a drive signal to a medical device, where the delivered power is desired to be at least 10 W and preferably between 10 W and 200 W; the delivered voltage is desired to be at least 20 V Rms  and preferably between 30 V Rms  and 120 V Rms ; the delivered current is designed to be at least 0.5 A RMS  and preferably between 1 A RMS  and 2 A RMS ; and the drive frequency for ultrasonic operation is desired to be at least 20 kHz and preferably between 30 kHz and 80 kHz; and the drive frequency for RF operation is desired to be at least 100 kHz and preferably between 250 kHz and 1 MHz. 
     In the above embodiment, the resonant circuits  43 - 1  and  43 - 2  were formed from capacitor-inductor-inductor elements. As those skilled in the art will appreciate, other resonant circuit designs with multiple capacitors and inductors in various series and parallel configurations or simpler LC resonant circuits may also be used. Also, in some applications there is no need for a transformer to step-up the drive voltage, as the FETs can deliver the required drive voltage. 
       FIG. 1  illustrates one way in which the batteries and the control electronics can be mounted within the handle of the medical device. As those skilled in the art will appreciate, the form factor of the handle may take many different designs. Indeed, it is not essential for the device to be battery powered, although this is preferred for some applications to avoid the need for power cords and the like. 
     The embodiment described above included a description of various novel features, including the novel ability to selectively apply ultrasonic energy or RF energy to the tissue gripped by the forceps, the novel way in which the microprocessor controlled the operation of the device in the electrical mode of operation; the way in which load current/voltage is measured and the way in which the batteries are protected using an active fuse circuit. As those skilled in the art will appreciate, these novel features do not need to be employed together. For example, the current/voltage sensing techniques described above can be used with other devices as can the active fuse circuit. Similarly, the way in which the electrical mode of operation is controlled by tracking the maximum power delivery condition and by using pulse skipping techniques can be used in a device that does not have an ultrasonic transducer. 
     In the above embodiment, an exemplary control algorithm for performing the cutting/cauterisation of the vessel or tissue gripped by the forceps was described. As those skilled in the art will appreciate, various different procedures may be used and the reader is referred to the literature describing the operation of such cutting/cauterisation devices for further details. 
     In the above embodiment, four FET switches were used to convert the DC voltage provided by the batteries into an alternating signal at the desired frequency. As those skilled in the art will appreciate, it is not necessary to use four switches—two switches may be used instead (using a half bridge circuit). Additionally, although FET switches were used, other switching devices, such as bipolar transistor switches may be used instead. However, MOSFETs are preferred due to their superior performance in terms of low losses when operating at the above described frequencies and current levels. 
     In the above embodiment, the I &amp; Q sampling circuitry  81  oversampled the sensed voltage/current signal in the ultrasonic mode of operation and undersampled the sensed voltage/current signal in the electrical mode of operation. As those skilled in the art will appreciate, this is not essential. Because of the synchronous nature of the sampling, samples may be taken more than once per period or once every n th  period if desired. The sampling rate used in the above embodiment was chosen to maximise the rate at which measurements were made available to the power controller  85  and the medical device control module  89  as this allows for better control of the applied power during the cauterisation process. 
     In the above embodiment, a 14V DC supply was provided. In other embodiments, lower (or higher) DC voltage sources may be provided. In this case, a larger (or smaller) transformer turns ratio may be provided to increase the load voltage to the desired level or lower operating voltages may be used. 
     In the above embodiment, the medical device was arranged to deliver a desired power (in the form of ultrasonic energy or electrical energy) to the tissue/vessel gripped by the forceps. In an alternative embodiment, the device may be arranged to deliver a desired current or a desired voltage level to the ultrasonic transducer or the forceps. 
     In the above embodiment the battery is shown integral to the medical device. In an alternative embodiment the battery may be packaged so as to clip on a belt on the surgeon or simply be placed on the Mayo stand. In this embodiment a relatively small two conductor cable would connect the battery pack to the medical device. 
     In the above embodiment, a microprocessor based control circuitry was provided. This is preferred due to the ease with which the microprocessor can be programmed to perform the above control actions using appropriate computer software. Such software can be provided on a tangible carrier, such as a CD-ROM or the like. Alternatively, hardware control circuitry can be used in place of the microprocessor based circuitry described above. 
     In the above embodiment, the user controlled whether the energy delivered to the vessel/tissue was ultrasonic energy or RF electrical energy. In alternative embodiments, the microprocessor may control the selection based on an internally generated control signal or in response to a control signal received from another device. 
     In the above embodiment, the active fuse circuit opened a switch that disconnected the bridge signal generator from the batteries. In an alternative embodiment, the active fuse circuit could instead switch in a large impedance between the batteries and the bridge signal generator to limit the current drawn from the batteries. Also, the switch could be used to disconnect the positive voltage supply from the signal generator instead of disconnecting the negative terminal of the batteries.