Patent Publication Number: US-7912145-B2

Title: Filter for a modulator and methods thereof

Description:
BACKGROUND OF THE INVENTION 
     In polar modulation, a signal is separated into its instantaneous amplitude and phase/frequency components (rather than into the classical in-phase (I) and quadrature (Q) components), and the amplitude component and phase/frequency component are modulated independently. The amplitude component may be modulated with any suitable amplitude modulation (AM) technique, while the phase/frequency component may be modulated using an analog phase locked loop (PLL). 
     To allow reasonable operation, the bandwidth of the PLL may be quite small, much smaller than the actual bandwidth of the transmission signal&#39;s instantaneous phase/frequency. For example, in the case where the PLL is fed by a sigma-delta converter that has a high pass noise nature, the loop filter may be narrow enough to attenuate the sigma-delta quantization noise and the phase noise of the PLL. A pre-emphasis filter may emphasize, prior to modulation, those frequency components that would be attenuated by the PLL. The pre-emphasis filter may employ inverse filtering to the linearized response of the PLL. This inverse filtering may yield a high-order infinite impulse response (IIR), which may suffer from stability problems. 
     Conventional practice involves calibration mechanisms in order to accurately calibrate the PLL to the predefined pre-emphasis filter. Without calibration, there is a risk that the pre-emphasis filter will not match the inverse to the PLL closed loop transfer function, which may result in enhancement of the phase noise. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Embodiments of the invention are illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like reference numerals indicate corresponding, analogous or similar elements, and in which: 
         FIG. 1  is a block-diagram illustration of an exemplary communication system including a transmitter according to some embodiments of the invention; 
         FIG. 2  is a block-diagram illustration of an exemplary fractional-N sigma-delta modulator according to some embodiments of the invention; 
         FIG. 3  is a flowchart illustration of an exemplary method to determine a transfer function of a filter, according to some embodiments of the invention; 
         FIG. 4  is a block-diagram illustration of an exemplary fractional-N sigma-delta modulator including an adaptive filter, according to some embodiments of the invention; 
         FIG. 5  is a block-diagram illustration of an exemplary linearized approximation to the exemplary fractional-N sigma-delta modulator of  FIG. 2 , according to some embodiments of the invention; 
         FIG. 6  is a block diagram of an exemplary system including an equalizer. 
     
    
    
     It will be appreciated that for simplicity and clarity of illustration, elements shown in the figures have not necessarily been drawn to scale. For example, the dimensions of some of the elements may be exaggerated relative to other elements for clarity. 
     DETAILED DESCRIPTION OF THE INVENTION 
     In the following detailed description, numerous specific details are set forth in order to provide a thorough understanding of embodiments of the invention. However it will be understood by those of ordinary skill in the art that embodiments of the invention may be practiced without these specific details. In other instances, well-known methods, procedures, components and circuits have not been described in detail so as not to obscure the description of embodiments of the invention. 
       FIG. 1  is a block-diagram illustration of an exemplary communication system including a transmitter according to an embodiment of the invention. A communication device  100  is able to communicate with a communication device  102  over a communication channel  104 . Although the scope of the invention is not limited in this respect, the communication system shown in  FIG. 1  may be part of a cellular communication system (with one of communication devices  100 ,  102  being a base station and the other a mobile station, or with both communication devices  100 ,  102  being mobile stations), a pager communication system, a personal digital assistant and a server, etc. A non-exhaustive list of examples for the communication system shown in  FIG. 1  includes a Global System for Mobile Communications (GSM) system, a General Packet Radio Service (GPRS) system, an Enhanced Data for GSM Evolution (EDGE) system, code division multiple access (CDMA), CDMA2000, Wideband-CDMA (WCDMA), AMPS and any other wireless standard. 
     Communication device  100  may include at least a transmitter  106  and an antenna  108 . Communication device  102  may include at least a receiver  110  and an antenna  112 . Antennas  108  and  112  may be of any desired kind such as, but not limited to, dipole, Yagi and multi-pole and the like. Moreover, communication device  100  may include a receiver (not shown). Similarly, communication device  102  may include a transmitter (not shown). 
     Transmitter  106  may include at least a baseband symbol generator  114  to generate a signal of baseband symbols, a splitter  116  to split the signal into its instantaneous amplitude and phase/frequency components, an amplitude path  118  to modulate and amplify the amplitude components, a fractional-N sigma-delta modulator  120  to modulate and up-convert the phase/frequency components, and a power amplifier  122  to amplify the output of fractional-N sigma-delta modulator  120  with a gain controlled by the output of amplitude path  118 . Antenna  108  is coupled to power amplifier  122  to transmit the output of power amplifier  122 . 
     Baseband symbol generator  114  may be implemented in accordance with a wireless standard. Splitter  116  may be implemented in hardware, software or firmware or any combination thereof. 
       FIG. 2  is a block-diagram illustration of an exemplary fractional-N sigma-delta modulator according to an embodiment of the invention, such as, for example, fractional-N sigma-delta modulator  120  of  FIG. 1 . Fractional-N sigma-delta modulator  120  has a digital portion and an analog portion. The digital portion may include at least a differentiator  202 , a filter  204  (such as, for example, a pre-emphasis filter), an amplifier  206 , a summer  208 , and a sigma-delta converter  210 . The analog portion, which implements an analog phase locked loop (PLL)  212 , may include at least a non-linear frequency divider  214 , a reference oscillator  216  to produce a signal having a frequency F REF , a non-linear phase-frequency detector  218 , a loop filter  220 , an amplifier  222  having a gain K V , and a voltage-controlled oscillator (VCO)  224 . The input of VCO  224  is denoted Y. PLL  212  is also known as a “fractional-N phase locked loop” unit. Amplifier  222  and VCO  224  may be implemented as a single unit (VCO  223 ); the separation into two elements is for the sake of analysis. 
     Reference oscillator  216  may produce a signal having a reference frequency F REF . An example of reference frequency F REF  is approximately 26 MHz, although other reference frequencies may be used instead. 
     Differentiator  202  may differentiate the phase symbols received from splitter  116  of  FIG. 1  to obtain the instantaneous frequency W of the baseband signal. Differentiator  202  may be implemented in hardware, software or firmware or any combination thereof. 
     Filter  204  may be determined so that the instantaneous frequency W is transferred to the input Y of VCO  224  with little or no distortion, as will be described in more detail hereinbelow. Any implementation of a digital filter is suitable for filter  204 . 
     Amplifier  206  may normalize the output of filter  204 , denoted X, to PLL reference frequency units. 
     Summer  208  may add the output of amplifier  206  with a number N+β. Both the integer number N and the non-integer number β having a value between 0 and 1 may be set according to an instruction from a base station regarding the average output frequency of a transmitter of a mobile station. For example, if reference frequency F REF  is approximately 26 MHz, then N may have a value in the range of 29-32 so that N·F REF  has a value of approximately 800 MHz. 
     Sigma-delta converter  210  may convert the output of summer  208  into an integer number that represents the instantaneous frequency division ratio of the PLL. 
     Non-linear frequency divider  214  may divide the output of VCO  224  by the integer number provided by sigma-delta converter  210  to produce a divided-frequency signal. 
     Non-linear phase-frequency detector  218  may compare the divided-frequency signal to the reference frequency signal produced by reference oscillator  216 . Non-linear phase-frequency detector  218  may produce a control signal that corresponds to the phase difference and/or frequency difference between the two signals. Any implementation of a phase-frequency detector is suitable for phase-frequency detector  218 . 
     The control signal, after smoothing by loop filter  220  and amplification by amplifier  222 , may be applied to VCO  224  so that VCO  224  synthesizes a modulated output carrier signal. 
     As mentioned above, filter  204  may be determined so that the instantaneous frequency W is transferred to the input Y of VCO  224  with little or no distortion (e.g. the overall response from W to Y is close to flat in the frequencies of interest). 
       FIG. 3  is a flowchart illustration of an exemplary method to determine a transfer function C of filter  204 , according to an embodiment of the invention. The method may include the following:
     a) building a linear model for PLL  212  and calculating the transfer function H(w) from the output X of filter  204  to the input Y to VCO  224  (block  302 );   b) optionally adding various impairments of the different PLL components (e.g. phase noises, variations of the PLL parameters from their nominal values, and the like) to the model (block  304 )—the variations of PLL parameters from their nominal values, such as, for example, capacitors, resistors, open loop gain, etc., may be due to production inaccuracies;   c) deciding on a topology for the transfer function C of filter  204  (block  306 ), and   d) calculating a transfer function C to minimize a predefined cost function related to the instantaneous frequency W and the input Y to VCO  224  (block  308 ).   

     In some embodiments, the transfer function C is a stable transfer function. 
     Mathematically, the transfer function C may be expressed as follows:
 
 C= ArgMin C {Cost( W,Y )}.
 
     In one embodiment, the cost function may be the mean square error (MSE) between the instantaneous frequency W and the input Y to VCO  224 , as follows: Cost(W,Y)=E{|W(t)−Y(t)| 2 }, where t is a time variable and E is an expectation operator. In yet another embodiment, the expectation operator is replaced by time averaging and the cost function becomes: 
               Cost   ⁢           ⁢     (     W   ,   Y     )       =       1   T     ⁢       ∫       -   T     /   2       T   /   2       ⁢                W   ⁡     (   t   )       -     Y   ⁡     (   t   )              2     ⁢     ⅆ   t                 
where T is a design parameter determining the integration time window.
 
     In another embodiment, the cost function may be a weighted MSE in the frequency domain, as follows: 
                 Cost   ⁢           ⁢     (     W   ,   Y     )       =     E   ⁢     {       ∫     -   ∞       +   ∞       ⁢       P   ⁡     (   w   )       ⁢              W   ⁡     (   w   )       -     Y   ⁡     (   w   )              2     ⁢     ⅆ   w         }         ,         
where P(w) is a user-defined, positive, weight function. For example, weight function P(w) may give more weight to those frequencies where spectral cleanliness is more important.
 
     In other embodiments, other cost functions may be used, such as, for example, a cost function that measures the spectral cleanliness of the overall transmission signal. 
     In the event that an MSE or weighted MSE cost function is used, then block  304  of the method may be redundant if all impairments may be represented as additive noise terms, since different choices of transfer function C will not affect that total contribution of these additive impairments to the MSE or weighted MSE cost. 
     Certain cost functions may be minimized using equalization theory, as will be described hereinbelow for a particular example. 
     Any suitable topology for transfer function C may be used in block  306 . For example, transfer function C may be a finite impulse response (FIR) of order p (in which case stability is guaranteed), or an infinite impulse response (IIR) having a rational transfer function of orders (p,q), etc. 
       FIG. 4  is a block diagram of an exemplary fractional-N sigma-delta modulator  420  including an adaptive filter, according to some embodiments of the invention. Fractional-N sigma-delta modulator  420  is similar to fractional-N sigma-delta modulator  120  of  FIG. 2 , and therefore similar components are referenced with the same reference numerals and will not be described in further detail. Fractional-N sigma-delta modulator  420  may include at least an adaptive filter  404  (such as, for example, an adaptive pre-emphasis filter). Adaptive filter  404  may include at least filter  204 , an analog-to-digital (A/D) converter  402  and an adaptive algorithm  403 . 
     Adaptive algorithm  403  compares the input to filter  204  (the instantaneous frequency W) to the input to VCO  223  (after digitization). Adaptive algorithm  403  adapts filter  204  according to the comparison. For example, if the error between instantaneous frequency W and input Y is defined by a particular cost function, then adaptive algorithm  403  may reduce the error iteratively. A non-exhaustive list of examples of adaptive mechanisms include Least Mean Squares (LMS), Recursive Least Squares (RLS), and the like. Thus, any impairments, offsets, drifts, etc. of the analog portion of the PLL may drive the filter values to those values that minimize a pre-specified adaptive mechanism cost function such as, for example, a mean squared error cost function. Variations in the PLL, such as for example, variations in temperature, voltage, aging, etc., may be compensated for by the adaptive algorithm. 
     Since adaptive algorithm  403  enables the digital values of filter  204  to be adapted to variations in the analog PLL, it may not be necessary to calibrate PLL  212  to predefined values for filter  204 . 
     Block  302  of the method of  FIG. 3  includes building a linear model of a PLL.  FIG. 5  is a block diagram of an exemplary linearized approximation to the exemplary fractional-N sigma-delta modulator of  FIG. 2 . In this approximation, which is made in the phase domain, PLL  212  has been approximated by a linearized PLL  512 , and sigma-delta converter  210  has been approximated by a linearized sigma-delta converter  510  including an all pass filter and an additive noise n(Σ−Δ). 
     Non-linear frequency divider  214  has been approximated by a linearized frequency divider  514  including an amplifier  511  having a gain of 2πF REF , a differentiator block  530  having a transfer function of s, where s is a Laplace domain variable, a subtraction block  532 , an amplifier  534  having a gain of 1/(N+β), and an integrator block  536  having a transfer function of 1/s. 
     VCO  224  has been approximated by a block  524  having a transfer function of 1/s. Non-linear phase-frequency detector  218  has been approximated by a subtraction block  518 , and loop filter  220  has been approximated by a block  520  having a transfer function T(s). 
     The transfer function from the output X of filter  204  to input Y to block  524  is then given by the following expression: 
     
       
         
           
             
               
                 
                   
                     
                       Y 
                       ⁡ 
                       
                         ( 
                         s 
                         ) 
                       
                     
                     
                       X 
                       ⁡ 
                       
                         ( 
                         s 
                         ) 
                       
                     
                   
                   = 
                   
                     
                       
                         ( 
                         
                           
                             K 
                             V 
                           
                           / 
                           
                             ( 
                             
                               N 
                               + 
                               β 
                             
                             ) 
                           
                         
                         ) 
                       
                       · 
                       
                         
                           T 
                           ⁡ 
                           
                             ( 
                             s 
                             ) 
                           
                         
                         / 
                         s 
                       
                     
                     
                       1 
                       + 
                       
                         
                           ( 
                           
                             
                               K 
                               V 
                             
                             / 
                             
                               ( 
                               
                                 N 
                                 + 
                                 β 
                               
                               ) 
                             
                           
                           ) 
                         
                         · 
                         
                           
                             T 
                             ⁡ 
                             
                               ( 
                               s 
                               ) 
                             
                           
                           / 
                           s 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     1 
                   
                   ) 
                 
               
             
           
         
       
     
     The design of the transfer function C of filter  204  may then be performed as follows. In order that the instantaneous frequency W be transferred to the input Y of VCO  224  with little or no frequency and phase distortion, the overall transfer function from W to Y may be of substantially 0 dB gain up to a cut-off frequency f 0  and may have substantially linear phase up to the cut-off frequency f 0 . 
     The cut-off frequency f 0  may be defined experimentally by observing the spectrum of the instantaneous frequency W signal. Alternatively, the cut-off frequency f 0  may be defined as narrow as possible while still meeting the requirements of a communication standard. Other definitions of the cut-off frequency f 0  are also within the scope of the invention. 
     One straightforward solution is to define the transfer function C from W to X up to the cut-off frequency f 0  as the inverse IIR to the transfer function Y(s)/X(s), as given by the following expression: 
                               X   ⁡     (   s   )         W   ⁡     (   s   )         =       1   +       (       K   V     /     (     N   +   β     )       )     ·       T   ⁡     (   s   )       /   s             (       K   V     /     (     N   +   β     )       )     ·       T   ⁡     (   s   )       /   s           ,                   s   =       j   ·   2     ⁢   π   ⁢           ⁢   f       ,     f   &lt;     f   0         ⁢                         (     Equation   ⁢           ⁢   2     )               
where j is the square root of −1.
 
     However, if the transfer function C were implemented as the inverse IIR to the transfer function Y(s)/X(s), then the following problems might arise:
     a) zeros and poles may need to be added to the inverse IIR to stabilize the filter;   b) additional poles or filters may need to be added to the inverse IIR to attenuate the frequencies above the cut-off frequency f 0  so that the sigma-delta converter will not be forced into saturation and so as not to increase the quantization noise; and   c) the order of the transfer function C would no longer be a design parameter; rather, it would have a one-to-one relation to the order of the closed loop transfer function. Therefore, according to some embodiments of the present invention, the transfer function C is not the inverse IIR to the transfer function Y(s)/X(s). Rather, the transfer function C of the filter is calculated to optimize predefined optimization criteria, as described hereinabove with respect to  FIG. 3 .   

     Block  302  of the method of  FIG. 3  includes calculating the transfer function C to minimize a predefined cost function related to the instantaneous frequency W and the input Y to the VCO of the PLL. The transfer function C may be calculated in a manner similar to the calculation of a minimum mean squared error (MMSE) equalizer, as will be explained in greater detail.  FIG. 6  is a block diagram of an exemplary system including an equalizer. 
     A known digital input signal u(i) may be subjected to an impulse response h, where the impulse response h is defined as the bi-linear transform of the transfer function Y(s)/X(s). 
     An additive white Gaussian noise n_AWGN(i) may be shaped by a shaping filter g. Noise shaping filter g may be a high pass filter with a cut-off above frequency f 0 , so that the equalizer response at high frequencies will be attenuated. 
     The combination of shaped noise n(i) and the output of impulse response h is denoted v(i). The digital signal v(i) may be subjected to a FIR filter C to yield a digital signal û(i). 
     FIR filter C may be calculated analytically to minimize the mean square error that is defined as
 
e≡u(i)−û(i).  (Equation 3)
 
     The following expressions may be useful in the calculation of FIR filter C:
 
{circumflex over ( u )}( i )=   C     H   ·  V ,   (Equation 4a)
 
   V =H·Ū+  N ,   (Equation 4b)
 
                     H   ≡       [           h   ⁡     (     M   +   L     )           ⋯         h   ⁡     (   L   )           ⋯         h   ⁡     (       -   M     +   L     )               ⋮                   ⋮                   ⋮             h   ⁡     (   M   )                         h   ⁡     (   0   )                         h   ⁡     (     -   M     )               ⋮                   ⋮                   ⋮             h   ⁡     (     M   -   L     )           ⋯         h   ⁡     (     -   L     )           ⋯         h   ⁡     (       -   M     -   L     )             ]       [       (       2   ⁢   L     +   1     )     ×     (       2   ⁢   M     +   1     )       ]         ,           (     Equation   ⁢           ⁢   4   ⁢   c     )                   U   _     ≡       [           u   ⁡     (     i   -   M     )               ⋮             u   ⁡     (   i   )                                 u   ⁡     (     i   +   M     )             ]       [       (       2   ⁢   M     +   1     )     ×   1     ]         ,           (     Equation   ⁢           ⁢   4   ⁢   d     )                   N   _     ≡       [           n   ⁡     (     i   +   L     )               ⋮             n   ⁡     (   i   )                                 u   ⁡     (     i   -   L     )             ]       [       (       2   ⁢   L     +   1     )     ×   1     ]         ,           (     Equation   ⁢           ⁢   4   ⁢   e     )               
where the bar denotes vector notation, the superscript H denotes the conjugate transpose, 2M+1 is assumed to be the length (i.e. number of coefficients) of impulse response h, and 2L+1 is assumed to be the length (i.e. number of coefficients) of the equalizer.
 
     The following assumptions may be made:
     1. There is no correlation between the error and the observations. Thus, E{e(i)·  V   H }=  0   H , where, again, E is an expectation operator.   2. There is no correlation between the input signal and the shaped noise. Thus E{e(i)·n(i+k)*}=0, ∀k, where the asterisk denotes the complex conjugate.   3. The input signal is independently distributed (ID). Thus   

     
       
         
           
             
               E 
               ⁢ 
               
                 { 
                 
                   
                     u 
                     ⁡ 
                     
                       ( 
                       i 
                       ) 
                     
                   
                   · 
                   
                     
                       u 
                       ⁡ 
                       
                         ( 
                         
                           i 
                           + 
                           k 
                         
                         ) 
                       
                     
                     * 
                   
                 
                 } 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         
                           
                             σ 
                             U 
                             2 
                           
                           ⁡ 
                           
                             ( 
                             i 
                             ) 
                           
                         
                         , 
                         
                           k 
                           = 
                           0 
                         
                       
                     
                   
                   
                     
                       
                         
                           0 
                           , 
                           
                             k 
                             ≠ 
                             0 
                           
                         
                         ⁢ 
                         
                             
                         
                       
                     
                   
                 
                 , 
                 
                   
                     where 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     
                       
                         σ 
                         U 
                         2 
                       
                       ⁡ 
                       
                         ( 
                         i 
                         ) 
                       
                     
                   
                   = 
                   
                     E 
                     ⁢ 
                     
                       
                         { 
                         
                           · 
                           
                             
                                
                               
                                 u 
                                 ⁡ 
                                 
                                   ( 
                                   i 
                                   ) 
                                 
                               
                                
                             
                             2 
                           
                         
                         } 
                       
                       . 
                     
                   
                 
               
             
           
         
       
         
         4. n_AWGN is additive white Gaussian noise. Thus 
       
    
     
       
         
           
             
               E 
               ⁢ 
               
                 { 
                 
                   n_AWGN 
                   ⁢ 
                   
                     
                       ( 
                       i 
                       ) 
                     
                     · 
                     n_AWGN 
                   
                   ⁢ 
                   
                     
                       ( 
                       
                         i 
                         + 
                         k 
                       
                       ) 
                     
                     * 
                   
                 
                 } 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         
                           σ 
                           N_AWGN 
                           2 
                         
                         , 
                         
                           k 
                           = 
                           0 
                         
                       
                     
                   
                   
                     
                       
                         
                           0 
                           , 
                           
                             k 
                             ≠ 
                             0 
                           
                         
                         ⁢ 
                         
                             
                         
                       
                     
                   
                 
                 , 
               
             
           
         
       
         
          where σ N     —     AWGN   2 =E{·|n 13  AWGN(i)| 2 } is a fixed value for all i. 
       
    
     From assumption 1 it follows that
 
 E{u ( i )·   V     H   }=E{û ( i )·   V     H }.  (Equation 5)
 
     By substituting Equations 4a, 4b, 4d and 4e into Equation 5, it follows that: 
                           E   ⁢     {         u   ^     ⁡     (   i   )       ·       V   _     H       }       =       ⁢           C   _     H     ·   E     ⁢     {       V   _     ·       V   _     H       }                   =       ⁢           C   _     H     ·   E     ⁢     {       (       H   ·     U   _       +     N   _       )     ·     (           U   _     H     ·     H   H       +       N   _     H       )       }                     =       ⁢         C   _     H     ·     [       E   ⁢     {     H   ·     U   _     ·       U   _     H     ·     H   H       }       +     E   ⁢     {       N   _     ·       N   _     H       }         ]         ,               =       ⁢         C   _     H     ·     [       H   ·     H   H     ·     σ   U   2       +     G   ·     G   H     ·     σ   N_AWGN   2         ]                     (     Equation   ⁢           ⁢   6     )               
where for simplicity, σ U   2 (i) is denoted by σ U   2 , and where matrix G is defined in terms of shaping filter g in a similar manner to the definition of matrix H in terms of impulse response h.
 
     Similarly, using Equation 4b and assumptions 2 and 3, it follows that: 
     
       
         
           
             
               
                 
                   
                     
                       
                         
                           
                             E 
                             ⁢ 
                             
                               { 
                               
                                 
                                   u 
                                   ⁡ 
                                   
                                     ( 
                                     i 
                                     ) 
                                   
                                 
                                 · 
                                 
                                   
                                     V 
                                     _ 
                                   
                                   H 
                                 
                               
                               } 
                             
                           
                           = 
                             
                           ⁢ 
                           
                             E 
                             ⁢ 
                             
                               { 
                               
                                 
                                   u 
                                   ⁡ 
                                   
                                     ( 
                                     i 
                                     ) 
                                   
                                 
                                 · 
                                 
                                   ( 
                                   
                                     
                                       H 
                                       · 
                                       
                                         U 
                                         _ 
                                       
                                     
                                     + 
                                     
                                       N 
                                       _ 
                                     
                                   
                                   ) 
                                 
                               
                               } 
                             
                           
                         
                       
                     
                     
                       
                         
                           = 
                             
                           ⁢ 
                           
                             
                               σ 
                               U 
                               2 
                             
                             · 
                             
                               [ 
                               
                                 
                                   h 
                                   ⁡ 
                                   
                                     ( 
                                     L 
                                     ) 
                                   
                                 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 … 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   h 
                                   ⁡ 
                                   
                                     ( 
                                     0 
                                     ) 
                                   
                                 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 … 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 
                                   h 
                                   ⁡ 
                                   
                                     ( 
                                     
                                       - 
                                       L 
                                     
                                     ) 
                                   
                                 
                               
                               ] 
                             
                           
                         
                       
                     
                   
                   . 
                 
               
               
                 
                   ( 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     7 
                   
                   ) 
                 
               
             
           
         
       
     
     From Equations 5, 6 and 7, and applying the complex conjugate operator, the MMSE analytical calculation of the FIR filter C is as follows: 
     
       
         
           
             
               
                 
                   C 
                   = 
                   
                     
                       
                         [ 
                         
                           
                             H 
                             · 
                             
                               H 
                               H 
                             
                             · 
                             
                               σ 
                               U 
                               2 
                             
                           
                           + 
                           
                             G 
                             · 
                             
                               G 
                               H 
                             
                             · 
                             
                               σ 
                               N_AWGN 
                               2 
                             
                           
                         
                         ] 
                       
                       
                         - 
                         1 
                       
                     
                     · 
                     
                       [ 
                       
                         
                           
                             
                               h 
                               ⁡ 
                               
                                 ( 
                                 L 
                                 ) 
                               
                             
                           
                         
                         
                           
                             ⋮ 
                           
                         
                         
                           
                             
                               h 
                               ⁡ 
                               
                                 ( 
                                 0 
                                 ) 
                               
                             
                           
                         
                         
                           
                             ⋮ 
                           
                         
                         
                           
                             
                               h 
                               ⁡ 
                               
                                 ( 
                                 
                                   - 
                                   L 
                                 
                                 ) 
                               
                             
                           
                         
                       
                       ] 
                     
                     · 
                     
                       
                         σ 
                         U 
                         2 
                       
                       . 
                     
                   
                 
               
               
                 
                   ( 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     8 
                   
                   ) 
                 
               
             
           
         
       
     
     FIR filter C may be calculated empirically. For example, a simulation environment may be built in which u(i) is the instantaneous frequency at the baseband, h(i) is the total response from the input to the sigma-delta converter until the input to the VCO, shaping filter g(i) is chosen so that shaped noise n(i) will be a high pass noise a cutoff frequency of which is beyond the band of interest of the total response. A simulation may then be run to determine empirical values for v(i). The following quantities may then be calculated: 
     
       
         
           
             
               
                 
                   
                     
                       R 
                       VV 
                     
                     = 
                     
                       
                         1 
                         L 
                       
                       · 
                       
                         
                           ∑ 
                           
                             k 
                             = 
                             0 
                           
                           
                             L 
                             - 
                             1 
                           
                         
                         ⁢ 
                         
                           
                             
                               V 
                               _ 
                             
                             ⁡ 
                             
                               ( 
                               
                                 i 
                                 - 
                                 k 
                               
                               ) 
                             
                           
                           · 
                           
                             
                               
                                 V 
                                 _ 
                               
                               H 
                             
                             ⁡ 
                             
                               ( 
                               
                                 i 
                                 - 
                                 k 
                               
                               ) 
                             
                           
                         
                       
                     
                   
                   , 
                   and 
                 
               
               
                 
                   ( 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     9 
                     ⁢ 
                     a 
                   
                   ) 
                 
               
             
             
               
                 
                   
                     R 
                     VU 
                   
                   = 
                   
                     
                       1 
                       L 
                     
                     · 
                     
                       
                         ∑ 
                         
                           k 
                           = 
                           0 
                         
                         
                           L 
                           - 
                           1 
                         
                       
                       ⁢ 
                       
                         
                           
                             V 
                             _ 
                           
                           ⁡ 
                           
                             ( 
                             
                               i 
                               - 
                               k 
                             
                             ) 
                           
                         
                         · 
                         
                           
                             
                               u 
                               * 
                             
                             ⁡ 
                             
                               ( 
                               
                                 i 
                                 - 
                                 k 
                               
                               ) 
                             
                           
                           . 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     9 
                     ⁢ 
                     b 
                   
                   ) 
                 
               
             
           
         
       
     
     The empirically calculated FIR MMSE filter is then given by:
 
   C =R   VV   −1   ·  R     VU .  (Equation 9c)
 
While certain features of the invention have been illustrated and described herein, many modifications, substitutions, changes, and equivalents will now occur to those of ordinary skill in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the spirit of the invention.