Patent Publication Number: US-9893710-B2

Title: High quality factor interconnect for RF circuits

Description:
RELATED APPLICATIONS 
     The present application claims priority to U.S. Provisional Patent Application No. 61/831,666, filed Jun. 6, 2013; U.S. Provisional Patent Application No. 61/860,932, filed Aug. 1, 2013; U.S. Provisional Patent Application No. 61/909,028, filed Nov. 26, 2013; U.S. Provisional Patent Application No. 61/938,884, filed Feb. 12, 2014; U.S. Provisional Patent Application No. 61/949,581, filed Mar. 7, 2014; U.S. Provisional Patent Application No. 61/951,844, filed Mar. 12, 2014; U.S. Provisional Patent Application No. 61/982,946, filed Apr. 23, 2014; U.S. Provisional Patent Application No. 61/982,952, filed Apr. 23, 2014; U.S. Provisional Patent Application No. 61/982,971, filed Apr. 23, 2014; and U.S. Provisional Patent Application No. 62/008,192, filed Jun. 5, 2014. 
     The present application is related to U.S. patent application Ser. No. 14/298,829, now U.S. Pat. No. 9,455,680, entitled “TUNABLE RF FILTER STRUCTURE FORMED BY A MATRIX OF WEAKLY COUPLED RESONATORS;” U.S. patent application Ser. No. 14/298,830, now U.S. Pat. No. 9,419,578 entitled “TUNABLE RF FILTER PATHS FOR TUNABLE RF FILTER STRUCTURES;” U.S. patent application Ser. No. 14/298,872, now U.S. Pat. No. 9,484,879, entitled “NONLINEAR CAPACITANCE LINEARIZATION;” U.S. patent application Ser. No. 14/298,863, filed on Jun. 6, 2014, entitled “TUNABLE RF FILTER BASED RF COMMUNICATIONS SYSTEM;” and U.S. patent application Ser. No. 14/298,852, now U.S. Pat. No. 9,614,490 entitled “MULTI-BAND INTERFERENCE OPTIMIZATION.” 
     All of the applications listed above are hereby incorporated herein by reference in their entireties. 
    
    
     FIELD OF THE DISCLOSURE 
     Embodiments of the present disclosure relate to radio frequency (RF) communications systems, which may include RF front-end circuitry, RF transceiver circuitry, RF amplifiers, direct current (DC)-DC converters, RF filters, RF antennas, RF switches, RF combiners, RF splitters, the like, or any combination thereof. 
     BACKGROUND 
     As wireless communications technologies evolve, wireless communications systems become increasingly sophisticated. As such, wireless communications protocols continue to expand and change to take advantage of the technological evolution. As a result, to maximize flexibility, many wireless communications devices must be capable of supporting any number of wireless communications protocols, each of which may have certain performance requirements, such as specific out-of-band emissions requirements, linearity requirements, or the like. Further, portable wireless communications devices are typically battery powered and need to be relatively small, and have low cost. As such, to minimize size, cost, and power consumption, RF circuitry in such a device needs to be as simple, small, flexible, and efficient as is practical. Thus, there is a need for RF circuitry in a communications device that is low cost, small, simple, flexible, and efficient. 
     SUMMARY 
     Embodiments of radio frequency (RF) devices are disclosed having interconnection paths with capacitive structures having improved quality (Q) factors. In one embodiment, an RF device includes an inductor having an inductor terminal and a semiconductor die. The semiconductor die includes one or more active semiconductor devices that include a device contact. The device contact provided by the one or more active semiconductor devices is positioned so as to be vertically aligned directly below the inductor terminal. The inductor terminal and the device contact are electrically connected with an interconnection path that includes a capacitive structure. To prevent or reduce current crowding, the interconnection path is vertically aligned so as to extend directly between the inductor terminal and the device contact. In this manner, the interconnection path electrically connects the inductor terminal and the device contact without degrading the Q factor of the RF device. 
     Those skilled in the art will appreciate the scope of the present disclosure and realize additional aspects thereof after reading the following detailed description of the preferred embodiments in association with the accompanying drawing figures. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings incorporated in and forming a part of this specification illustrate several aspects of the disclosure, and together with the description serve to explain the principles of the disclosure. 
         FIG. 1  shows traditional communications circuitry according to the prior art. 
         FIG. 2  shows the traditional communications circuitry according to the prior art. 
         FIG. 3  shows the traditional communications circuitry according to the prior art. 
         FIG. 4  shows RF communications circuitry according to one embodiment of the RF communications circuitry. 
         FIG. 5  is a graph illustrating filtering characteristics of a first tunable RF filter path and a second tunable RF filter path illustrated in  FIG. 4  according to one embodiment of the first tunable RF filter path and the second tunable RF filter path. 
         FIGS. 6A and 6B  are graphs illustrating filtering characteristics of the first tunable RF filter path and the second tunable RF filter path, respectively, illustrated in  FIG. 4  according to an alternate embodiment of the first tunable RF filter path and the second tunable RF filter path, respectively. 
         FIG. 7  shows the RF communications circuitry according to one embodiment of the RF communications circuitry. 
         FIG. 8  shows the RF communications circuitry according to an alternate embodiment of the RF communications circuitry. 
         FIGS. 9A and 9B  are graphs illustrating filtering characteristics of the first tunable RF filter path and the second tunable RF filter path, respectively, illustrated in  FIG. 8  according to an additional embodiment of the first tunable RF filter path and the second tunable RF filter path. 
         FIGS. 10A and 10B  are graphs illustrating filtering characteristics of a first traditional RF duplexer and a second traditional RF duplexer, respectively, illustrated in  FIG. 3  according to the prior art. 
         FIG. 11  shows the RF communications circuitry according to one embodiment of the RF communications circuitry. 
         FIG. 12  shows the RF communications circuitry according to an alternate embodiment of the RF communications circuitry. 
         FIG. 13  shows the RF communications circuitry according to an additional embodiment of the RF communications circuitry. 
         FIG. 14  shows the RF communications circuitry according to another embodiment of the RF communications circuitry. 
         FIG. 15  shows the RF communications circuitry according to a further embodiment of the RF communications circuitry. 
         FIG. 16  shows the RF communications circuitry according to one embodiment of the RF communications circuitry. 
         FIG. 17  shows the RF communications circuitry according to an alternate embodiment of the RF communications circuitry. 
         FIG. 18  shows the RF communications circuitry according to an additional embodiment of the RF communications circuitry. 
         FIG. 19  shows the RF communications circuitry according to another embodiment of the RF communications circuitry. 
         FIG. 20  shows the RF communications circuitry according to a further embodiment of the RF communications circuitry. 
         FIG. 21  illustrates one embodiment of a tunable radio frequency (RF) filter structure that defines multiple tunable RF filtering paths that are independent of each other. 
         FIG. 22  illustrates one embodiment of a tunable RF filter path shown in  FIG. 21  having cross-coupling capacitors arranged in a V-bridge structure. 
         FIG. 23  illustrates another embodiment of the tunable RF filter path shown in  FIG. 21  having cross-coupling capacitors arranged in an X-bridge structure. 
         FIG. 24  illustrates another embodiment of the tunable RF filter path shown in  FIG. 21  having a cross-coupling capacitor arranged in a single positive bridge structure. 
         FIG. 25  illustrates another embodiment of the tunable RF filter path shown in  FIG. 21  having cross-coupling capacitors arranged in an H-bridge structure. 
         FIG. 26  illustrates another embodiment of the tunable RF filter path shown in  FIG. 21  having cross-coupling capacitors arranged in a double H-bridge structure. 
         FIG. 27  illustrates another embodiment of the tunable RF filter path shown in  FIG. 21  having four weakly coupled resonators with magnetic and electric couplings between them. 
         FIGS. 28A-28D  disclose different embodiments of a tunable RF filter structure, each with a different number of input terminals and output terminals. 
         FIG. 29  illustrates one embodiment of a tunable radio frequency (RF) filter structure having four resonators and cross-coupling capacitive structures electrically connected between the four resonators so as to form a 2×2 matrix with the four resonators. In alternative embodiments, fewer (e.g., three) resonators or more (e.g., five or more) resonators may be provided. 
         FIG. 30  illustrates another embodiment of a tunable RF filter structure having M number of rows and N number of columns of resonators that are electrically connected by cross-coupling capacitive structures so that the tunable RF filter structure is arranged so as to form an M×N two-dimensional matrix of the resonators. 
         FIG. 31  illustrates the tunable RF filter structure shown in  FIG. 30  electrically connected to various RF antennas. 
         FIG. 32  illustrates the tunable RF filter structure shown in  FIG. 30  with two tunable RF filter paths highlighted for performing Multiple Input Multiple Output (MIMO), Single Input Multiple Output (SIMO), Multiple Input Single Output (MISO), and Single Input Single Output (SISO) operations. 
         FIG. 33  illustrates another embodiment of a tunable RF filter structure with amplifier stages electrically connected within and between tunable RF filter paths. 
         FIG. 34  illustrates an embodiment of a tunable RF filter structure integrated into an integrated circuit (IC) package with multiple and separate semiconductor dies. 
         FIG. 35  illustrates an embodiment of the same tunable RF filter structure shown in  FIG. 34 , but now integrated into an IC package with a single semiconductor die. 
         FIG. 36  illustrates one embodiment of a tunable RF filter structure having resonators and cross-coupling capacitive structures electrically connected between the resonators so as to form a three-dimensional matrix of the resonators. 
         FIG. 37  illustrates an embodiment of a radio frequency (RF) device integrated into an integrated circuit (IC) package, wherein the RF device includes an inductor having inductor terminals electrically connected to device contacts of active semiconductor devices with interconnection paths that may include capacitive structures. 
         FIG. 38  illustrates another embodiment of the RF device having an inductor terminal electrically connected to a device contact of an active semiconductor device with an interconnection path having a capacitive structure, where the active semiconductor device is grounded to a virtual or a real ground connection. 
         FIG. 39  illustrates an embodiment of RF devices that are electrically connected with a trace formed from a metallic layer on a package board or alternatively to a thick top metal layer of a Back-End-Of-Line (BEOL). 
         FIG. 40  illustrates an embodiment of RF devices that are electrically connected with a trace within a BEOL). 
         FIG. 41  illustrates a top view of an embodiment of an interconnection path having a metal-insulator-metal (MIM) capacitive structure formed within the interconnection path. 
         FIG. 42  illustrates a top view of an embodiment of an interconnection path having a metal-oxide-metal (MOM) capacitive structure. 
         FIG. 43  illustrates a top view of an embodiment of an interconnection path having metal-to-metal (MTM) capacitive structures formed by parallel metallic walls formed across drain contacts provided by active semiconductor devices. 
         FIG. 44  illustrates a top view of an embodiment of an interconnection path having MTM capacitive structures formed by posts surrounded by metallic cages formed across drain contacts provided by active semiconductor devices. 
         FIG. 45  illustrates another embodiment of an MTM capacitive structure formed using parallel metallic walls formed across a drain contact and a source contact of an active semiconductor device. 
         FIG. 46  illustrates another embodiment of an MTM capacitive structure formed using parallel metallic walls formed across a drain contact, a source contact, and a gate contact of an active semiconductor device. 
     
    
    
     DETAILED DESCRIPTION 
     The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the disclosure and illustrate the best mode of practicing the disclosure. Upon reading the following description in light of the accompanying drawings, those skilled in the art will understand the concepts of the disclosure and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims. 
     RF communications circuitry, which includes a first RF filter structure, is disclosed according to a first embodiment of the present disclosure. The first RF filter structure includes a first tunable RF filter path and a second tunable RF filter path. The first tunable RF filter path includes a pair of weakly coupled resonators. Additionally, a first filter parameter of the first tunable RF filter path is tuned based on a first filter control signal. A first filter parameter of the second tunable RF filter path is tuned based on a second filter control signal. 
     In one embodiment of the first RF filter structure, the first tunable RF filter path is directly coupled between a first common connection node and a first connection node. The second tunable RF filter path is directly coupled between a second connection node and the first common connection node. 
     In one embodiment of the RF communications system, the first tunable RF filter path and the second tunable RF filter path do not significantly load one another at frequencies of interest. As such, by directly coupling the first tunable RF filter path and the second tunable RF filter path to the first common connection node; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications system. In one embodiment of the RF communications system, the first common connection node is coupled to an antenna. 
     Embodiments of the RF communications system include frequency division duplex (FDD) applications, time division duplex (TDD) applications, carrier-aggregation (CA) applications, multiple antenna applications, MIMO applications, hybrid applications, applications supporting multiple communications bands, the like, or any combination thereof. 
       FIG. 1  shows traditional communications circuitry  10  according to the prior art. The traditional communications circuitry  10  illustrated in  FIG. 1  is a time-division duplex (TDD) system, which is capable of transmitting and receiving RF signals, but not simultaneously. Such a system may also be called a half-duplex system. Additionally, the traditional communications circuitry  10  may be used as a simplex system, which is a system that only transmits RF signals or only receives RF signals. Traditional communications systems often use fixed frequency filters. As a result, to cover multiple communications bands, switching elements are needed to select between different signal paths. 
     The traditional communications circuitry  10  includes traditional RF system control circuitry  12 , traditional RF front-end circuitry  14 , and a first RF antenna  16 . The traditional RF front-end circuitry  14  includes traditional RF front-end control circuitry  18 , first traditional antenna matching circuitry  20 , first traditional RF receive circuitry  22 , first traditional RF transmit circuitry  24 , a first traditional RF switch  26 , and a second traditional RF switch  28 . The first traditional RF switch  26  is coupled between the first traditional antenna matching circuitry  20  and the first traditional RF receive circuitry  22 . The second traditional RF switch  28  is coupled between the first traditional antenna matching circuitry  20  and the first traditional RF transmit circuitry  24 . The first RF antenna  16  is coupled to the first traditional antenna matching circuitry  20 . The first traditional antenna matching circuitry  20  provides at least partial impedance matching between the first RF antenna  16  and either the first traditional RF receive circuitry  22  or the first traditional RF transmit circuitry  24 . 
     The traditional RF system control circuitry  12  provides the necessary control functions needed to facilitate RF communications between the traditional communications circuitry  10  and other RF devices. The traditional RF system control circuitry  12  processes baseband signals needed for the RF communications. As such, the traditional RF system control circuitry  12  provides a first traditional upstream transmit signal TUT 1  to the first traditional RF transmit circuitry  24 . The first traditional upstream transmit signal TUT 1  may be a baseband transmit signal, an intermediate frequency (IF) transmit signal, or an RF transmit signal. Conversely, the traditional RF system control circuitry  12  receives a first traditional downstream receive signal TDR 1  from the first traditional RF receive circuitry  22 . The first traditional downstream receive signal TDR 1  may be a baseband receive signal, an IF receive signal, or an RF receive signal. 
     The first traditional RF transmit circuitry  24  may include up-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. Similarly, the first traditional RF receive circuitry  22  may include down-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. 
     The traditional RF system control circuitry  12  provides a traditional front-end control signal TFEC to the traditional RF front-end control circuitry  18 . The traditional RF front-end control circuitry  18  provides a first traditional switch control signal TCS 1  and a second traditional switch control signal TCS 2  to the first traditional RF switch  26  and the second traditional RF switch  28 , respectively, based on the traditional front-end control signal TFEC. As such, the traditional RF system control circuitry  12  controls the first traditional RF switch  26  and the second traditional RF switch  28  via the traditional front-end control signal TFEC. The first traditional RF switch  26  is in one of an ON state and an OFF state based on the first traditional switch control signal TCS 1 . The second traditional RF switch  28  is in one of an ON state and an OFF state based on the second traditional switch control signal TCS 2 . 
     Half-duplex operation of the traditional communications circuitry  10  is accomplished using the first traditional RF switch  26  and the second traditional RF switch  28 . When the traditional communications circuitry  10  is transmitting RF signals via the first RF antenna  16 , the first traditional RF switch  26  is in the OFF state and the second traditional RF switch  28  is in the ON state. As such, the first traditional antenna matching circuitry  20  is electrically isolated from the first traditional RF receive circuitry  22  and the first traditional antenna matching circuitry  20  is electrically coupled to the first traditional RF transmit circuitry  24 . In this regard, the traditional RF system control circuitry  12  provides the first traditional upstream transmit signal TUT 1  to the first traditional RF transmit circuitry  24 , which provides a traditional transmit signal TTX to the first RF antenna  16  via the second traditional RF switch  28  and the first traditional antenna matching circuitry  20  based on the first traditional upstream transmit signal TUT 1 . 
     When the traditional communications circuitry  10  is receiving RF signals via the first RF antenna  16 , the first traditional RF switch  26  is in the ON state and the second traditional RF switch  28  is in the OFF state. As such, the first traditional antenna matching circuitry  20  is isolated from the first traditional RF transmit circuitry  24  and the first traditional antenna matching circuitry  20  is electrically coupled to the first traditional RF receive circuitry  22 . In this regard, the first traditional antenna matching circuitry  20  receives the RF signals from the first RF antenna  16  and forwards the RF signals via the first traditional RF switch  26  to the first traditional RF receive circuitry  22 . The first traditional RF switch  26  provides a traditional receive signal TRX to the first traditional RF receive circuitry  22 , which provides a first traditional downstream receive signal TDR 1  to the traditional RF system control circuitry  12  based on the traditional receive signal TRX. 
     Since the traditional communications circuitry  10  illustrated in  FIG. 1  is a half-duplex system, during operation, the first traditional RF switch  26  and the second traditional RF switch  28  are not simultaneously in the ON state. Therefore, the first traditional RF receive circuitry  22  and the first traditional RF transmit circuitry  24  are isolated from one another. As such, the first traditional RF receive circuitry  22  and the first traditional RF transmit circuitry  24  are prevented from interfering with one another. 
       FIG. 2  shows the traditional communications circuitry  10  according to the prior art. The traditional communications circuitry  10  illustrated in  FIG. 2  is similar to the traditional communications circuitry  10  illustrated in  FIG. 1 , except in the traditional communications circuitry  10  illustrated in  FIG. 2 , the traditional RF front-end control circuitry  18 , the first traditional RF switch  26 , and the second traditional RF switch  28  are omitted, and the traditional RF front-end circuitry  14  further includes a first traditional RF duplexer  30 . The first traditional RF duplexer  30  is coupled between the first traditional antenna matching circuitry  20  and the first traditional RF receive circuitry  22 , and is further coupled between the first traditional antenna matching circuitry  20  and the first traditional RF transmit circuitry  24 . 
     The traditional communications circuitry  10  illustrated in  FIG. 2  may be used as a TDD system or a simplex system. However, the traditional communications circuitry  10  illustrated in  FIG. 2  may also be used as a frequency-division duplex (FDD) system, which is capable of transmitting and receiving RF signals simultaneously. Such a system may also be called a full-duplex system. 
     When the traditional communications circuitry  10  is transmitting RF signals via the first RF antenna  16 , the traditional RF system control circuitry  12  provides the first traditional upstream transmit signal TUT 1  to the first traditional RF transmit circuitry  24 , which provides the traditional transmit signal TTX to the first RF antenna  16  via first traditional RF duplexer  30  based on the first traditional upstream transmit signal TUT 1 . 
     When the traditional communications circuitry  10  is receiving RF signals via the first RF antenna  16 , the first traditional antenna matching circuitry  20  receives the RF signals from the first RF antenna  16  and forwards the RF signals via the first traditional RF duplexer  30  to the first traditional RF receive circuitry  22 . As such, the first traditional RF duplexer  30  provides the traditional receive signal TRX to the first traditional RF receive circuitry  22 , which provides the first traditional downstream receive signal TDR 1  to the traditional RF system control circuitry  12  based on the traditional receive signal TRX. 
     The first traditional RF duplexer  30  provides filtering, such that the first traditional RF receive circuitry  22  and the first traditional RF transmit circuitry  24  are substantially isolated from one another. As such, the first traditional RF receive circuitry  22  and the first traditional RF transmit circuitry  24  are prevented from interfering with one another. Traditional FDD systems using duplexers with high rejection ratios have a fixed frequency transfer. Covering multiple communications bands requires multiple duplexers and switches to route RF signals through appropriate signal paths. 
       FIG. 3  shows the traditional communications circuitry  10  according to the prior art. The traditional communications circuitry  10  illustrated in  FIG. 3  is a carrier aggregation (CA) based system, which is capable of transmitting or receiving multiple simultaneous transmit signals or multiple simultaneous receive signals, respectively, or both. Each of the simultaneous transmit signals is in a frequency band that is different from each frequency band of a balance of the simultaneous transmit signals. Similarly, each of the simultaneous receive signals is in a frequency band that is different from each frequency band of a balance of the simultaneous receive signals. The traditional communications circuitry  10  may operate as a simplex system, a half-duplex system, or a full-duplex system. 
     The traditional communications circuitry  10  includes the traditional RF system control circuitry  12 , the traditional RF front-end circuitry  14 , the first RF antenna  16 , and a second RF antenna  32 . The traditional RF front-end circuitry  14  includes the first traditional antenna matching circuitry  20 , the first traditional RF receive circuitry  22 , the first traditional RF transmit circuitry  24 , the first traditional RF duplexer  30 , first traditional antenna switching circuitry  34 , a second traditional RF duplexer  36 , a third traditional RF duplexer  38 , second traditional antenna matching circuitry  40 , second traditional antenna switching circuitry  42 , a fourth traditional RF duplexer  44 , a fifth traditional RF duplexer  46 , a sixth traditional RF duplexer  48 , second traditional RF receive circuitry  50 , and second traditional RF transmit circuitry  52 . Traditional CA systems use fixed frequency filters and diplexers, triplexers, or both to combine signal paths, which increases complexity. Alternatively, additional switch paths may be used, but may degrade performance. 
     The first traditional antenna matching circuitry  20  is coupled between the first RF antenna  16  and the first traditional antenna switching circuitry  34 . The second traditional antenna matching circuitry  40  is coupled between the second RF antenna  32  and the second traditional antenna switching circuitry  42 . The first traditional RF duplexer  30  is coupled between the first traditional antenna switching circuitry  34  and the first traditional RF receive circuitry  22 , and is further coupled between the first traditional antenna switching circuitry  34  and the first traditional RF transmit circuitry  24 . The second traditional RF duplexer  36  is coupled between the first traditional antenna switching circuitry  34  and the first traditional RF receive circuitry  22 , and is further coupled between the first traditional antenna switching circuitry  34  and the first traditional RF transmit circuitry  24 . The third traditional RF duplexer  38  is coupled between the first traditional antenna switching circuitry  34  and the first traditional RF receive circuitry  22 , and is further coupled between the first traditional antenna switching circuitry  34  and the first traditional RF transmit circuitry  24 . 
     The fourth traditional RF duplexer  44  is coupled between the second traditional antenna switching circuitry  42  and the second traditional RF receive circuitry  50 , and is further coupled between the second traditional antenna switching circuitry  42  and the second traditional RF transmit circuitry  52 . The fifth traditional RF duplexer  46  is coupled between the second traditional antenna switching circuitry  42  and the second traditional RF receive circuitry  50 , and is further coupled between the second traditional antenna switching circuitry  42  and the second traditional RF transmit circuitry  52 . The sixth traditional RF duplexer  48  is coupled between the second traditional antenna switching circuitry  42  and the second traditional RF receive circuitry  50 , and is further coupled between the second traditional antenna switching circuitry  42  and the second traditional RF transmit circuitry  52 . 
     The first traditional RF duplexer  30  is associated with a first aggregated receive band, a first aggregated transmit band, or both. The second traditional RF duplexer  36  is associated with a second aggregated receive band, a second aggregated transmit band, or both. The third traditional RF duplexer  38  is associated with a third aggregated receive band, a third aggregated transmit band, or both. The fourth traditional RF duplexer  44  is associated with a fourth aggregated receive band, a fourth aggregated transmit band, or both. The fifth traditional RF duplexer  46  is associated with a fifth aggregated receive band, a fifth aggregated transmit band, or both. The sixth traditional RF duplexer  48  is associated with a sixth aggregated receive band, a sixth aggregated transmit band, or both. 
     The first traditional antenna switching circuitry  34  couples a selected one of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38  to the first traditional antenna matching circuitry  20 . Therefore, the first RF antenna  16  is associated with a selected one of the first aggregated receive band, the second aggregated receive band, and the third aggregated receive band; with a selected one of the first aggregated transmit band, the second aggregated transmit band, and the third aggregated transmit band; or both. 
     Similarly, the second traditional antenna switching circuitry  42  couples a selected one of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48  to the second traditional antenna matching circuitry  40 . Therefore, the second RF antenna  32  is associated with a selected one of the fourth aggregated receive band, the fifth aggregated receive band, and the sixth aggregated receive band; with a selected one of the fourth aggregated transmit band, the fifth aggregated transmit band, and the sixth aggregated transmit band; or both. 
     During transmit CA, the traditional RF system control circuitry  12  provides the first traditional upstream transmit signal TUT 1  to the first traditional RF transmit circuitry  24 , which forwards the first traditional upstream transmit signal TUT 1  to the first RF antenna  16  for transmission via the selected one of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38 ; via the first traditional antenna switching circuitry  34 ; and via the first traditional antenna matching circuitry  20 . 
     Additionally, during transmit CA, the traditional RF system control circuitry  12  provides a second traditional upstream transmit signal TUT 2  to the second traditional RF transmit circuitry  52 , which forwards the second traditional upstream transmit signal TUT 2  to the second RF antenna  32  for transmission via the selected one of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48 ; via the second traditional antenna switching circuitry  42 ; and via the second traditional antenna matching circuitry  40 . 
     During receive CA, the first RF antenna  16  forwards a received RF signal to the first traditional RF receive circuitry  22  via the first traditional antenna matching circuitry  20 , the first traditional antenna switching circuitry  34 , and the selected one of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38 . The first traditional RF receive circuitry  22  provides the first traditional downstream receive signal TDR 1  to the traditional RF system control circuitry  12  based on the received RF signal. 
     Additionally, during receive CA, the second RF antenna  32  forwards a received RF signal to the second traditional RF receive circuitry  50  via the second traditional antenna matching circuitry  40 , the second traditional antenna switching circuitry  42 , and the selected one of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48 . The second traditional RF receive circuitry  50  provides a second traditional downstream receive signal TDR 2  to the traditional RF system control circuitry  12  based on the received RF signal. 
     Since only the selected one of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38  is coupled to the first traditional antenna matching circuitry  20 ; the first traditional antenna switching circuitry  34  isolates each of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38  from one another; and prevents each of the first traditional RF duplexer  30 , the second traditional RF duplexer  36 , and the third traditional RF duplexer  38  from interfering with one another. 
     Similarly, since only the selected one of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48  is coupled to the second traditional antenna matching circuitry  40 ; the second traditional antenna matching circuitry  40  isolates each of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48  from one another; and prevents each of the fourth traditional RF duplexer  44 , the fifth traditional RF duplexer  46 , and the sixth traditional RF duplexer  48  from interfering with one another. 
       FIG. 4  shows RF communications circuitry  54  according to one embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  includes RF system control circuitry  56 , RF front-end circuitry  58 , and the first RF antenna  16 . The RF front-end circuitry  58  includes a first RF filter structure  60 , RF receive circuitry  62 , and RF transmit circuitry  64 . The first RF filter structure  60  includes a first tunable RF filter path  66  and a second tunable RF filter path  68 . Additionally, the first RF filter structure  60  has a first connection node  70 , a second connection node  72 , and a first common connection node  74 . In one embodiment of the RF system control circuitry  56 , the RF system control circuitry  56  is an RF transceiver. In one embodiment of the first tunable RF filter path  66 , the first tunable RF filter path  66  includes a pair of weakly coupled resonators R( 1 , 1 ), R( 1 , 2 ) ( FIG. 22 ). As such, in one embodiment of the first RF filter structure  60 , the RF filter structure  60  includes the pair of weakly coupled resonators R( 1 , 1 ), R( 1 , 2 ) ( FIG. 21 ). 
     In alternate embodiments of the first RF filter structure  60 , any or all of the first connection node  70 , the second connection node  72 , and the first common connection node  74  are external to the first RF filter structure  60 . In one embodiment of the first tunable RF filter path  66 , the first tunable RF filter path  66  includes a first pair (not shown) of weakly coupled resonators. In one embodiment of the second tunable RF filter path  68 , the second tunable RF filter path  68  includes a second pair (not shown) of weakly coupled resonators. 
     In one embodiment of the first RF filter structure  60 , the first tunable RF filter path  66  is directly coupled between the first common connection node  74  and the first connection node  70 , the second tunable RF filter path  68  is directly coupled between the second connection node  72  and the first common connection node  74 , and the first RF antenna  16  is directly coupled to the first common connection node  74 . In another embodiment of the RF communications circuitry  54 , the first RF antenna  16  is omitted. Additionally, the RF receive circuitry  62  is coupled between the first connection node  70  and the RF system control circuitry  56 , and the RF transmit circuitry  64  is coupled between the second connection node  72  and the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  is a first RF receive filter, such that the first RF antenna  16  forwards a received RF signal via the first common connection node  74  to provide a first upstream RF receive signal RU 1  to the first tunable RF filter path  66 , which receives and filters the first upstream RF receive signal RU 1  to provide a first filtered RF receive signal RF 1  to the RF receive circuitry  62 . The RF receive circuitry  62  may include down-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. The RF receive circuitry  62  processes the first filtered RF receive signal RF 1  to provide a first receive signal RX 1  to the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the second tunable RF filter path  68  is a first RF transmit filter, such that the RF system control circuitry  56  provides a first transmit signal TX 1  to the RF transmit circuitry  64 , which processes the first transmit signal TX 1  to provide a first upstream RF transmit signal TU 1  to the second tunable RF filter path  68 . The RF transmit circuitry  64  may include up-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. The second tunable RF filter path  68  receives and filters the first upstream RF transmit signal TU 1  to provide a first filtered RF transmit signal TF 1 , which is transmitted via the first common connection node  74  by the first RF antenna  16 . 
     The RF system control circuitry  56  provides a first filter control signal FCS 1  to the first tunable RF filter path  66  and provides a second filter control signal FCS 2  to the second tunable RF filter path  68 . As such, in one embodiment of the RF communications circuitry  54 , the RF system control circuitry  56  tunes a first filter parameter of the first tunable RF filter path  66  using the first filter control signal FCS 1 . Additionally, the RF system control circuitry  56  tunes a first filter parameter of the second tunable RF filter path  68  using the second filter control signal FCS 2 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  and the second tunable RF filter path  68  do not significantly load one another at frequencies of interest. As such, by directly coupling the first tunable RF filter path  66  and the second tunable RF filter path  68  to the first common connection node  74 ; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications circuitry  54 . Since tunable RF filters can support multiple communications bands using a single signal path, they can simplify front-end architectures by eliminating switching and duplexing components. 
     In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is used as an FDD communications system, such that the first upstream RF receive signal RU 1  and the first filtered RF transmit signal TF 1  are full-duplex signals. In an alternate embodiments of the RF communications circuitry  54 , the RF communications circuitry  54  is used as a TDD communications system, such that the first upstream RF receive signal RU 1  and the first filtered RF transmit signal TF 1  are half-duplex signals. In additional embodiments of the RF communications circuitry  54 , the RF communications circuitry  54  is used as a simplex communications system, such that the first upstream RF receive signal RU 1  is a simplex signal and the first filtered RF transmit signal TF 1  is not present. In other embodiments of the RF communications circuitry  54 , the RF communications circuitry  54  is used as a simplex communications system, such that the first upstream RF receive signal RU 1  is not present and the first filtered RF transmit signal TF 1  is a simplex signal. 
       FIG. 5  is a graph illustrating filtering characteristics of the first tunable RF filter path  66  and the second tunable RF filter path  68  illustrated in  FIG. 4  according to one embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 . The first tunable RF filter path  66  is a first RF bandpass filter, which functions as the first RF receive filter, and the second tunable RF filter path  68  is a second RF bandpass filter, which functions as the first RF transmit filter. A bandwidth  76  of the first RF bandpass filter, a center frequency  78  of the first RF bandpass filter, a bandwidth  80  of the second RF bandpass filter, a center frequency  82  of the second RF bandpass filter, a frequency  84  of the first upstream RF receive signal RU 1  ( FIG. 4 ), and a frequency  86  of the first filtered RF transmit signal TF 1  ( FIG. 4 ) are shown. Operation of the first RF bandpass filter and the second RF bandpass filter is such that the first RF bandpass filter and the second RF bandpass filter do not significantly interfere with one another. In this regard, the bandwidth  76  of the first RF bandpass filter does not overlap the bandwidth  80  of the second RF bandpass filter. 
     In one embodiment of the first RF receive filter and the first RF transmit filter, the first RF receive filter and the first RF transmit filter in combination function as an RF duplexer. As such, a duplex frequency  88  of the RF duplexer is about equal to a difference between the frequency  84  of the first upstream RF receive signal RU 1  ( FIG. 4 ) and the frequency  86  of the first filtered RF transmit signal TF 1  ( FIG. 4 ). 
     In one embodiment of the first tunable RF filter path  66 , the first filter parameter of the first tunable RF filter path  66  is tunable based on the first filter control signal FCS 1 . In an alternate embodiment of the first tunable RF filter path  66 , both the first filter parameter of the first tunable RF filter path  66  and a second filter parameter of the first tunable RF filter path  66  are tunable based on the first filter control signal FCS 1 . Similarly, in one embodiment of the second tunable RF filter path  68 , the first filter parameter of the second tunable RF filter path  68  is tunable based on the second filter control signal FCS 2 . In an alternate embodiment of the second tunable RF filter path  68 , both the first filter parameter of the second tunable RF filter path  68  and a second filter parameter of the second tunable RF filter path  68  are tunable based on the second filter control signal FCS 2 . 
     The first filter parameter of the first tunable RF filter path  66  is the center frequency  78  of the first RF bandpass filter. The second filter parameter of the first tunable RF filter path  66  is the bandwidth  76  of the first RF bandpass filter. The first filter parameter of the second tunable RF filter path  68  is the center frequency  82  of the second RF bandpass filter. The second filter parameter of the second tunable RF filter path  68  is the bandwidth  80  of the second RF bandpass filter. 
       FIGS. 6A and 6B  are graphs illustrating filtering characteristics of the first tunable RF filter path  66  and the second tunable RF filter path  68 , respectively, illustrated in  FIG. 4  according to an alternate embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , respectively. The first tunable RF filter path  66  is an RF lowpass filter and the second tunable RF filter path  68  is an RF highpass filter.  FIG. 6A  shows a frequency response curve  90  of the RF lowpass filter and  FIG. 6B  shows a frequency response curve  92  of the RF highpass filter. Additionally  FIG. 6A  shows a break frequency  94  of the RF lowpass filter and  FIG. 6B  shows a break frequency  96  of the RF highpass filter. Both  FIGS. 6A and 6B  show the frequency  84  of the first upstream RF receive signal RU 1  ( FIG. 4 ), the frequency  86  of the first filtered RF transmit signal TF 1  ( FIG. 4 ), and the duplex frequency  88  of the RF duplexer for clarification. However, the RF lowpass filter and the RF highpass filter in combination function as an RF diplexer. The first filter parameter of the first tunable RF filter path  66  is the break frequency  94  of the RF lowpass filter. In one embodiment of the RF lowpass filter, the RF lowpass filter has bandpass filter characteristics. The first filter parameter of the second tunable RF filter path  68  is the break frequency  96  of the RF highpass filter. In one embodiment of the RF highpass filter, the RF highpass filter has bandpass filter characteristics. In one embodiment of the RF diplexer, the break frequency  96  of the RF highpass filter is about equal to the break frequency  94  of the RF lowpass filter. 
       FIG. 7  shows the RF communications circuitry  54  according to one embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 7  is similar to the RF communications circuitry  54  illustrated in  FIG. 4 , except in the RF front-end circuitry  58  illustrated in  FIG. 7 , the RF transmit circuitry  64  ( FIG. 4 ) is omitted and the RF front-end circuitry  58  further includes RF front-end control circuitry  98 . 
     The RF system control circuitry  56  provides a front-end control signal FEC to the RF front-end control circuitry  98 . The RF front-end control circuitry  98  provides the first filter control signal FCS 1  and the second filter control signal FCS 2  based on the front-end control signal FEC. In the RF communications circuitry  54  illustrated in  FIG. 4 , the RF system control circuitry  56  provides the first filter control signal FCS 1  and the second filter control signal FCS 2  directly. In general, the RF communications circuitry  54  includes control circuitry, which may be either the RF system control circuitry  56  or the RF front-end control circuitry  98 , that provides the first filter control signal FCS 1  and the second filter control signal FCS 2 . As such, in one embodiment of the RF communications circuitry  54 , the control circuitry tunes a first filter parameter of the first tunable RF filter path  66  using the first filter control signal FCS 1 . Additionally, the control circuitry tunes a first filter parameter of the second tunable RF filter path  68  using the second filter control signal FCS 2 . In an additional embodiment of the RF communications circuitry  54 , the control circuitry further tunes a second filter parameter of the first tunable RF filter path  66  using the first filter control signal FCS 1 ; and the control circuitry further tunes a second filter parameter of the second tunable RF filter path  68  using the second filter control signal FCS 2 . 
     In alternate embodiments of the first RF filter structure  60 , any or all of the first connection node  70 , the second connection node  72 , and the first common connection node  74  are external to the first RF filter structure  60 . In one embodiment of the first tunable RF filter path  66 , the first tunable RF filter path  66  includes a first pair (not shown) of weakly coupled resonators. In one embodiment of the second tunable RF filter path  68 , the second tunable RF filter path  68  includes a second pair (not shown) of weakly coupled resonators. 
     In one embodiment of the first RF filter structure  60 , the first tunable RF filter path  66  is directly coupled between the first common connection node  74  and the first connection node  70 , the second tunable RF filter path  68  is directly coupled between the second connection node  72  and the first common connection node  74 , and the first RF antenna  16  is directly coupled to the first common connection node  74 . In another embodiment of the RF communications circuitry  54 , the first RF antenna  16  is omitted. Additionally, the RF receive circuitry  62  is coupled between the first connection node  70  and the RF system control circuitry  56 , and the RF receive circuitry  62  is further coupled between the second connection node  72  and the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  is a first RF receive filter, such that the first RF antenna  16  forwards a first received RF signal via the first common connection node  74  to provide a first upstream RF receive signal RU 1  to the first tunable RF filter path  66 , which receives and filters the first upstream RF receive signal RU 1  to provide a first filtered RF receive signal RF 1  to the RF receive circuitry  62 . Additionally, the second tunable RF filter path  68  is a second RF receive filter, such that the first RF antenna  16  forwards a second received RF signal via the first common connection node  74  to provide a second upstream RF receive signal RU 2  to the second tunable RF filter path  68 , which receives and filters the second upstream RF receive signal RU 2  to provide a second filtered RF receive signal RF 2  to the RF receive circuitry  62 . 
     The RF receive circuitry  62  may include down-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. The RF receive circuitry  62  processes the first filtered RF receive signal RF 1  to provide a first receive signal RX 1  to the RF system control circuitry  56 . Additionally, the RF receive circuitry  62  processes the second filtered RF receive signal RF 2  to provide a second receive signal RX 2  to the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  and the second tunable RF filter path  68  do not significantly load one another at frequencies of interest. As such, by directly coupling the first tunable RF filter path  66  and the second tunable RF filter path  68  to the first common connection node  74 ; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications circuitry  54 . 
     In this regard, in one embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter having a unique center frequency. As such, the first filter parameter of each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a unique center frequency. 
     In an alternate embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a lowpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a highpass filter. As such, the first filter parameter of each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In an additional embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a lowpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter. As such, the first filter parameter of one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a center frequency, and the first filter parameter of another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In an additional embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a highpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter. As such, the first filter parameter of one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a center frequency, and the first filter parameter of another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a receive only CA system, such that the first tunable RF filter path  66 , which is the first RF receive filter, and the second tunable RF filter path  68 , which is the second RF receive filter, simultaneously receive and filter the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2 , respectively, via the first common connection node  74 . As such, the first RF filter structure  60  functions as a de-multiplexer. In this regard, each of the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  has a unique carrier frequency. Using receive CA may increase an effective receive bandwidth of the RF communications circuitry  54 . 
     In another embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a receive only communications system, such that the first tunable RF filter path  66 , which is the first RF receive filter, and the second tunable RF filter path  68 , which is the second RF receive filter, do not simultaneously receive and filter the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2 , respectively. As such, the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  are nonsimultaneous signals. Each of the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  may be associated with a unique RF communications band. 
       FIG. 8  shows the RF communications circuitry  54  according to an alternate embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 8  is similar to the RF communications circuitry  54  illustrated in  FIG. 7 , except in the RF front-end circuitry  58  illustrated in  FIG. 8 , the RF receive circuitry  62  is omitted and the RF transmit circuitry  64  is included. 
     The RF system control circuitry  56  provides the front-end control signal FEC to the RF front-end control circuitry  98 . The RF front-end control circuitry  98  provides the first filter control signal FCS 1  and the second filter control signal FCS 2  based on the front-end control signal FEC. In the RF communications circuitry  54  illustrated in  FIG. 4 , the RF system control circuitry  56  provides the first filter control signal FCS 1  and the second filter control signal FCS 2  directly. In general, the RF communications circuitry  54  includes control circuitry, which may be either the RF system control circuitry  56  or the RF front-end control circuitry  98 , that provides the first filter control signal FCS 1  and the second filter control signal FCS 2 . As such, in one embodiment of the RF communications circuitry  54 , the control circuitry tunes a first filter parameter of the first tunable RF filter path  66  using the first filter control signal FCS 1 . Additionally, the control circuitry tunes a first filter parameter of the second tunable RF filter path  68  using the second filter control signal FCS 2 . In an additional embodiment of the RF communications circuitry  54 , the control circuitry further tunes a second filter parameter of the first tunable RF filter path  66  using the first filter control signal FCS 1 ; and the control circuitry further tunes a second filter parameter of the second tunable RF filter path  68  using the second filter control signal FCS 2 . 
     In alternate embodiments of the first RF filter structure  60 , any or all of the first connection node  70 , the second connection node  72 , and the first common connection node  74  are external to the first RF filter structure  60 . In one embodiment of the first tunable RF filter path  66 , the first tunable RF filter path  66  includes a first pair (not shown) of weakly coupled resonators. In one embodiment of the second tunable RF filter path  68 , the second tunable RF filter path  68  includes a second pair (not shown) of weakly coupled resonators. 
     In one embodiment of the first RF filter structure  60 , the first tunable RF filter path  66  is directly coupled between the first common connection node  74  and the first connection node  70 , the second tunable RF filter path  68  is directly coupled between the second connection node  72  and the first common connection node  74 , and the first RF antenna  16  is directly coupled to the first common connection node  74 . In another embodiment of the RF communications circuitry  54 , the first RF antenna  16  is omitted. Additionally, the RF transmit circuitry  64  is coupled between the first connection node  70  and the RF system control circuitry  56 , and the RF transmit circuitry  64  is further coupled between the second connection node  72  and the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  is a first RF transmit filter, such that the RF system control circuitry  56  provides the first transmit signal TX 1  to the RF transmit circuitry  64 , which processes the first transmit signal TX 1  to provide a first upstream RF transmit signal TU 1  to the first tunable RF filter path  66 . Similarly, the second tunable RF filter path  68  is a second RF transmit filter, such that the RF system control circuitry  56  provides a second transmit signal TX 2  to the RF transmit circuitry  64 , which processes the second transmit signal TX 2  to provide a second upstream RF transmit signal TU 2  to the second tunable RF filter path  68 . 
     The RF transmit circuitry  64  may include up-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. The first tunable RF filter path  66  receives and filters the first upstream RF transmit signal TU 1  to provide the first filtered RF transmit signal TF 1 , which is transmitted via the first common connection node  74  by the first RF antenna  16 . Similarly, the second tunable RF filter path  68  receives and filters the second upstream RF transmit signal TU 2  to provide a second filtered RF transmit signal TF 2 , which is transmitted via the first common connection node  74  by the first RF antenna  16 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66  and the second tunable RF filter path  68  do not significantly load one another at frequencies of interest. As such, by directly coupling the first tunable RF filter path  66  and the second tunable RF filter path  68  to the first common connection node  74 ; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications circuitry  54 . 
     In this regard, in one embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter having a unique center frequency. As such, the first filter parameter of each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a unique center frequency. 
     In an alternate embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a lowpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a highpass filter. As such, the first filter parameter of each of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In an additional embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a lowpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter. As such, the first filter parameter of one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a center frequency, and the first filter parameter of another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In an additional embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a highpass filter, and another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a bandpass filter. As such, the first filter parameter of one of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a center frequency, and the first filter parameter of another of the first tunable RF filter path  66  and the second tunable RF filter path  68  is a break frequency. 
     In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a transmit only CA system, such that the first tunable RF filter path  66 , which is the first RF transmit filter, and the second tunable RF filter path  68 , which is the second RF transmit filter, simultaneously receive and filter the first upstream RF transmit signal TU 1  and the second upstream RF transmit signal TU 2 , respectively, to simultaneously provide the first filtered RF transmit signal TF 1  and the second filtered RF transmit signal TF 2 , respectively, via the first common connection node  74 . As such, the first RF filter structure  60  functions as a multiplexer. In this regard, each of the first filtered RF transmit signal TF 1  and the second filtered RF transmit signal TF 2  has a unique carrier frequency. Using transmit CA may increase an effective transmit bandwidth of the RF communications circuitry  54 . 
     In another embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a transmit only communications system, such that the first tunable RF filter path  66 , which is the first RF transmit filter, and the second tunable RF filter path  68 , which is the second RF transmit filter, do not simultaneously receive and filter the first upstream RF transmit signal TU 1  and the second upstream RF transmit signal TU 2 , respectively. As such, the first filtered RF transmit signal TF 1  and the second filtered RF transmit signal TF 2  are nonsimultaneous signals. Each of the first filtered RF transmit signal TF 1  and the second filtered RF transmit signal TF 2  may be associated with a unique RF communications band. 
       FIGS. 9A and 9B  are graphs illustrating filtering characteristics of the first tunable RF filter path  66  and the second tunable RF filter path  68 , respectively, illustrated in  FIG. 8  according to an additional embodiment of the first tunable RF filter path  66  and the second tunable RF filter path  68 , respectively.  FIG. 9A  shows a frequency response curve  100  of the first tunable RF filter path  66  and  FIG. 9B  shows a frequency response curve  102  of the second tunable RF filter path  68 . The first tunable RF filter path  66  and the second tunable RF filter path  68  are both bandpass filters having the frequency response curves  100 ,  102  illustrated in  FIGS. 9A and 9B , respectively. In this regard, the first tunable RF filter path  66  and the second tunable RF filter path  68  can be directly coupled to one another via the first common connection node  74  ( FIG. 8 ) without interfering with one another. 
       FIGS. 10A and 10B  are graphs illustrating filtering characteristics of the first traditional RF duplexer  30  and the second traditional RF duplexer  36 , respectively, illustrated in  FIG. 3  according to the prior art.  FIG. 10A  shows a frequency response curve  104  of the first traditional RF duplexer  30  and  FIG. 10B  shows a frequency response curve  106  of the second traditional RF duplexer  36 . There is interference  108  between the frequency response curve  104  of the first traditional RF duplexer  30  and the frequency response curve  106  of the second traditional RF duplexer  36  as shown in  FIGS. 10A and 10B . In this regard, the first traditional RF duplexer  30  and the second traditional RF duplexer  36  cannot be directly coupled to one another without interfering with one another. To avoid interference between different filters, traditional systems use RF switches to disconnect unused filters. 
       FIG. 11  shows the RF communications circuitry  54  according to one embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 11  is similar to the RF communications circuitry  54  illustrated in  FIG. 8 , except in the RF communications circuitry  54  illustrated in  FIG. 11 , the RF front-end circuitry  58  further includes the RF receive circuitry  62  and the first RF filter structure  60  further includes a third tunable RF filter path  110  and a fourth tunable RF filter path  112 . Additionally, the RF front-end circuitry  58  has the first connection node  70 , the second connection node  72 , the first common connection node  74 , a third connection node  114  and a fourth connection node  116 , such that all of the first connection node  70 , the second connection node  72 , the first common connection node  74 , the third connection node  114  and the fourth connection node  116  are external to the first RF filter structure  60 . In an alternate of the RF front-end circuitry  58 , any or all of the first connection node  70 , the second connection node  72 , the first common connection node  74 , a third connection node  114  and a fourth connection node  116  are internal to the first RF filter structure  60 . 
     The RF front-end control circuitry  98  further provides a third filter control signal FCS 3  to the third tunable RF filter path  110  and a fourth filter control signal FCS 4  to the fourth tunable RF filter path  112  based on the front-end control signal FEC. In one embodiment of the RF communications circuitry  54 , the control circuitry tunes a first filter parameter of the third tunable RF filter path  110  using the third filter control signal FCS 3 . Additionally, the control circuitry tunes a first filter parameter of the fourth tunable RF filter path  112  using the fourth filter control signal FCS 4 . In an additional embodiment of the RF communications circuitry  54 , the control circuitry further tunes a second filter parameter of the third tunable RF filter path  110  using the third filter control signal FCS 3 ; and the control circuitry further tunes a second filter parameter of the fourth tunable RF filter path  112  using the fourth filter control signal FCS 4 . 
     In one embodiment of the third tunable RF filter path  110 , the third tunable RF filter path  110  includes a third pair (not shown) of weakly coupled resonators. In one embodiment of the fourth tunable RF filter path  112 , the fourth tunable RF filter path  112  includes a fourth pair (not shown) of weakly coupled resonators. 
     In one embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , the third tunable RF filter path  110  is directly coupled between the first common connection node  74  and the third connection node  114 , and the fourth tunable RF filter path  112  is directly coupled between the fourth connection node  116  and the first common connection node  74 . In another embodiment of the RF communications circuitry  54 , the first RF antenna  16  is omitted. Additionally, the RF receive circuitry  62  is coupled between the third connection node  114  and the RF system control circuitry  56 , and the RF receive circuitry  62  is further coupled between the fourth connection node  116  and the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the third tunable RF filter path  110  is the first RF receive filter, such that the first RF antenna  16  forwards a first received RF signal via the first common connection node  74  to provide the first upstream RF receive signal RU 1  to the third tunable RF filter path  110 , which receives and filters the first upstream RF receive signal RU 1  to provide the first filtered RF receive signal RF 1  to the RF receive circuitry  62 . Additionally, the fourth tunable RF filter path  112  is a second RF receive filter, such that the first RF antenna  16  forwards a second received RF signal via the first common connection node  74  to provide the second upstream RF receive signal RU 2  to the fourth tunable RF filter path  112 , which receives and filters the second upstream RF receive signal RU 2  to provide the second filtered RF receive signal RF 2  to the RF receive circuitry  62 . 
     The RF receive circuitry  62  may include down-conversion circuitry, amplification circuitry, power supply circuitry, filtering circuitry, switching circuitry, combining circuitry, splitting circuitry, dividing circuitry, clocking circuitry, the like, or any combination thereof. The RF receive circuitry  62  processes the first filtered RF receive signal RF 1  to provide the first receive signal RX 1  to the RF system control circuitry  56 . Additionally, the RF receive circuitry  62  processes the second filtered RF receive signal RF 2  to provide the second receive signal RX 2  to the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , and the fourth tunable RF filter path  112  do not significantly load one another at frequencies of interest. As such, by directly coupling the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , and the fourth tunable RF filter path  112  to the first common connection node  74 ; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications circuitry  54 . 
     In this regard, in one embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , each of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a bandpass filter having a unique center frequency. As such, the first filter parameter of each of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a unique center frequency. 
     In an alternate embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , one of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a lowpass filter, and another of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a highpass filter. As such, the first filter parameter of each of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a break frequency. 
     In an additional embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , one of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a lowpass filter, and another of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a bandpass filter. As such, the first filter parameter of one of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a center frequency, and the first filter parameter of another of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a break frequency. 
     In an additional embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , one of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a highpass filter, and another of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a bandpass filter. As such, the first filter parameter of one of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a center frequency, and the first filter parameter of another of the third tunable RF filter path  110  and the fourth tunable RF filter path  112  is a break frequency. 
     In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a CA system, such that the third tunable RF filter path  110 , which is the first RF receive filter, and the fourth tunable RF filter path  112 , which is the second RF receive filter, simultaneously receive and filter the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2 , respectively, via the first common connection node  74 . As such, the first RF filter structure  60  functions as a de-multiplexer using the third tunable RF filter path  110  and the fourth tunable RF filter path  112 . In one embodiment of the first RF filter structure  60 , the first RF filter structure  60  further functions as a multiplexer using the first tunable RF filter path  66  and the second tunable RF filter path  68 . In this regard, each of the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  has a unique carrier frequency. 
     In another embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is a receive communications system, such that the third tunable RF filter path  110 , which is the first RF receive filter, and the fourth tunable RF filter path  112 , which is the second RF receive filter, do not simultaneously receive and filter the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2 , respectively. As such, the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  are nonsimultaneous signals. Each of the first upstream RF receive signal RU 1  and the second upstream RF receive signal RU 2  may be associated with a unique RF communications band. 
       FIG. 12  shows the RF communications circuitry  54  according to an alternate embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 12  is similar to the RF communications circuitry  54  illustrated in  FIG. 11 , except the RF communications circuitry  54  illustrated in  FIG. 12  further includes the second RF antenna  32 . Additionally, the RF front-end circuitry  58  further includes a second common connection node  118  and a second RF filter structure  120 . The third tunable RF filter path  110  and the fourth tunable RF filter path  112  are included in the second RF filter structure  120  instead of being included in the first RF filter structure  60 . Instead of being coupled to the first common connection node  74 , the third tunable RF filter path  110  and the fourth tunable RF filter path  112  are coupled to the second common connection node  118 . In one embodiment of the third tunable RF filter path  110  and the fourth tunable RF filter path  112 , the third tunable RF filter path  110  and the fourth tunable RF filter path  112  are directly coupled to the second common connection node  118 . In one embodiment of the RF communications circuitry  54 , the second RF antenna  32  is coupled to the second common connection node  118 . 
       FIG. 13  shows the RF communications circuitry  54  according to an additional embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 13  is similar to the RF communications circuitry  54  illustrated in  FIG. 12 , except in the RF communications circuitry  54  illustrated in  FIG. 13 , the RF front-end control circuitry  98  provides a front-end status signal FES to the RF system control circuitry  56 . Additionally, the RF front-end control circuitry  98  provides a first calibration control signal CCS 1  and up to and including an N TH  calibration control signal CCSN to the first RF filter structure  60 . The RF front-end control circuitry  98  provides a P TH  calibration control signal CCSP and up to and including an X TH  calibration control signal CCSX to the second RF filter structure  120 . Details of the first RF filter structure  60  and the second RF filter structure  120  are not shown to simplify  FIG. 13 . 
     The first RF filter structure  60  provides a first calibration status signal CSS 1  and up to and including a Q TH  calibration status signal CSSQ to the RF front-end control circuitry  98 . The second RF filter structure  120  provides an R TH  calibration status signal CSSR and up to and including a Y TH  calibration status signal CSSY to the RF front-end control circuitry  98 . In an alternate embodiment of the RF front-end circuitry  58 , any or all of the N TH  calibration control signal CCSN, the Q TH  calibration status signal CSSQ, the X TH  calibration control signal CCSX, and the Y TH  calibration status signal CSSY are omitted. 
     In one embodiment of the RF front-end circuitry  58 , the RF front-end circuitry  58  operates in one of a normal operating mode and a calibration mode. During the calibration mode, the RF front-end control circuitry  98  performs a calibration of the first RF filter structure  60 , the second RF filter structure  120 , or both. As such, the RF front-end control circuitry  98  provides any or all of the filter control signals FCS 1 , FCS 2 , FCS 3 , FCS 4  and any or all of the calibration control signals CCS 1 , CCSN, CCSP, CCSX needed for calibration. Further, the RF front-end control circuitry  98  receives any or all of the calibration status signals CSS 1 , CSSQ, CSSR, CSSY needed for calibration. 
     During the normal operating mode, the RF front-end control circuitry  98  provides any or all of the filter control signals FCS 1 , FCS 2 , FCS 3 , FCS 4  and any or all of the calibration control signals CCS 1 , CCSN, CCSP, CCSX needed for normal operation. Further, the RF front-end control circuitry  98  receives any or all of the calibration status signals CSS 1 , CSSQ, CSSR, CSSY needed for normal operation. Any or all of the calibration control signals CCS 1 , CCSN, CCSP, CCSX may be based on the front-end control signal FEC. The front-end status signal FES may be based on any or all of the calibration status signals CSS 1 , CSSQ, CSSR, CSSY. Further, during the normal operating mode, the RF front-end circuitry  58  processes signals as needed for normal operation. Other embodiments described in the present disclosure may be associated with normal operation. 
     The RF communications circuitry  54  illustrated in  FIG. 13  includes the first RF antenna  16  and the second RF antenna  32 . In general, the RF communications circuitry  54  is a multiple antenna system. A single-input single-output (SISO) antenna system is a system in which RF transmit signals may be transmitted from the first RF antenna  16  and RF receive signals may be received via the second RF antenna  32 . In one embodiment of the RF communications circuitry  54 , the antenna system in the RF communications circuitry  54  is a SISO antenna system, as illustrated in  FIG. 13 . 
     A single-input multiple-output (SIMO) antenna system is a system in which RF transmit signals may be simultaneously transmitted from the first RF antenna  16  and the second RF antenna  32 , and RF receive signals may be received via the second RF antenna  32 . In an alternate embodiment of the RF communications circuitry  54 , the second RF filter structure  120  is coupled to the RF transmit circuitry  64 , such that the antenna system in the RF communications circuitry  54  is a SIMO antenna system. 
     A multiple-input single-output (MISO) antenna system is a system in which RF transmit signals may be transmitted from the first RF antenna  16 , and RF receive signals may be simultaneously received via the first RF antenna  16  and the second RF antenna  32 . In an additional embodiment of the RF communications circuitry  54 , the first RF filter structure  60  is coupled to the RF receive circuitry  62 , such that the antenna system in the RF communications circuitry  54  is a MISO antenna system. 
     A multiple-input multiple-output (MIMO) antenna system is a system in which RF transmit signals may be simultaneously transmitted from the first RF antenna  16  and the second RF antenna  32 , and RF receive signals may be simultaneously received via the first RF antenna  16  and the second RF antenna  32 . In another embodiment of the RF communications circuitry  54 , the second RF filter structure  120  is coupled to the RF transmit circuitry  64  and the first RF filter structure  60  is coupled to the RF receive circuitry  62 , such that the antenna system in the RF communications circuitry  54  is a MIMO antenna system. 
       FIG. 14  shows the RF communications circuitry  54  according to another embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 14  is similar to the RF communications circuitry  54  illustrated in  FIG. 11 , except in the RF communications circuitry  54  illustrated in  FIG. 14 , the first RF filter structure  60  further includes a fifth tunable RF filter path  122  and a sixth tunable RF filter path  124 , and the RF front-end circuitry  58  further includes a fifth connection node  126  and a sixth connection node  128 . Additionally, the RF front-end control circuitry  98  shown in  FIG. 11  is not shown in  FIG. 14  to simplify  FIG. 14 . 
     In one embodiment of the fifth tunable RF filter path  122 , the fifth tunable RF filter path  122  includes a fifth pair (not shown) of weakly coupled resonators. In one embodiment of the sixth tunable RF filter path  124 , the sixth tunable RF filter path  124  includes a sixth pair (not shown) of weakly coupled resonators. 
     In one embodiment of the fifth tunable RF filter path  122  and the sixth tunable RF filter path  124 , the fifth tunable RF filter path  122  is directly coupled between the first common connection node  74  and the fifth connection node  126 , and the sixth tunable RF filter path  124  is directly coupled between the sixth connection node  128  and the first common connection node  74 . In another embodiment of the RF communications circuitry  54 , the first RF antenna  16  is omitted. Additionally, the RF receive circuitry  62  is further coupled between the sixth connection node  128  and the RF system control circuitry  56 , and the RF transmit circuitry  64  is further coupled between the fifth connection node  126  and the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the sixth tunable RF filter path  124  is a third RF receive filter, such that the first RF antenna  16  forwards a third received RF signal via the first common connection node  74  to provide a third upstream RF receive signal RU 3  to the sixth tunable RF filter path  124 , which receives and filters the third upstream RF receive signal RU 3  to provide a third filtered RF receive signal RF 3  to the RF receive circuitry  62 , which processes the third filtered RF receive signal RF 3  to provide the third receive signal RX 3  to the RF system control circuitry  56 . 
     In one embodiment of the RF communications circuitry  54 , the fifth tunable RF filter path  122  is a third RF transmit filter, such that the RF system control circuitry  56  provides a third transmit signal TX 3  to the RF transmit circuitry  64 , which processes the third transmit signal TX 3  to provide a third upstream RF transmit signal TU 3  to the fifth tunable RF filter path  122 . The fifth tunable RF filter path  122  receives and filters the third upstream RF transmit signal TU 3  to provide a third filtered RF transmit signal TF 3 , which is transmitted via the first common connection node  74  by the first RF antenna  16 . 
     In one embodiment of the RF communications circuitry  54 , the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  do not significantly load one another at frequencies of interest. Therefore, antenna switching circuitry  34 ,  42  ( FIG. 3 ) may be avoided. As such, by directly coupling the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122  and the sixth tunable RF filter path  124  to the first common connection node  74 ; front-end RF switching elements may be avoided, thereby reducing cost, size, and non-linearity; and increasing efficiency and flexibility of the RF communications circuitry  54 . 
     In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  is an FDD communications system, such that each of the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  is a bandpass filter having a unique center frequency. As such, in one embodiment of the RF system control circuitry  56 , the first filter parameter of each of the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  is a unique center frequency. 
       FIG. 15  shows the RF communications circuitry  54  according to a further embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 15  is similar to the RF communications circuitry  54  illustrated in  FIG. 4 , except in the RF communications circuitry  54  illustrated in  FIG. 15 , the RF front-end circuitry  58  further includes an RF antenna switch  130  and the third connection node  114 . Additionally, the first RF filter structure  60  further includes the third tunable RF filter path  110 . Instead of the first RF antenna  16  being directly coupled to the first common connection node  74 , as illustrated in  FIG. 4 , the RF antenna switch  130  is coupled between the first RF antenna  16  and the first common connection node  74 . As such, the first common connection node  74  is coupled to the first RF antenna  16  via the RF antenna switch  130 . In this regard, the RF communications circuitry  54  is a hybrid RF communications system. 
     The RF antenna switch  130  has an antenna switch common connection node  132 , an antenna switch first connection node  134 , an antenna switch second connection node  136 , and an antenna switch third connection node  138 . The antenna switch common connection node  132  is coupled to the first RF antenna  16 . In one embodiment of the RF antenna switch  130 , the antenna switch common connection node  132  is directly coupled to the first RF antenna  16 . The antenna switch first connection node  134  is coupled to the first common connection node  74 . In one embodiment of the RF antenna switch  130 , the antenna switch first connection node  134  is directly coupled to the first common connection node  74 . The antenna switch second connection node  136  may be coupled to other circuitry (not shown). The antenna switch third connection node  138  may be coupled to other circuitry (not shown). In another embodiment of the RF antenna switch  130 , the antenna switch third connection node  138  is omitted. In a further embodiment of the RF antenna switch  130 , the RF antenna switch  130  has at least one additional connection node. 
     The RF system control circuitry  56  provides a switch control signal SCS to the RF antenna switch  130 . As such, the RF system control circuitry  56  selects one of the antenna switch first connection node  134 , the antenna switch second connection node  136 , and the antenna switch third connection node  138  to be coupled to the antenna switch common connection node  132  using the switch control signal SCS. 
     The third tunable RF filter path  110  is directly coupled between the first common connection node  74  and the third connection node  114 . In one embodiment of the RF communications circuitry  54 , the third tunable RF filter path  110  is a second RF receive filter, such that the first RF antenna  16  forwards a received RF signal via the RF antenna switch  130  and the first common connection node  74  to provide the second upstream RF receive signal RU 2  to the third tunable RF filter path  110 , which receives and filters the second upstream RF receive signal RU 2  to provide the second filtered RF receive signal RF 2  to the RF receive circuitry  62 . The RF receive circuitry  62  processes the second filtered RF receive signal RF 2  to provide a second receive signal RX 2  to the RF system control circuitry  56 . 
     The RF system control circuitry  56  further provides the third filter control signal FCS 3 . As such, in one embodiment of the RF communications circuitry  54 , the RF system control circuitry  56  tunes a first filter parameter of the third tunable RF filter path  110  using the third filter control signal FCS 3 . In one embodiment of the RF communications circuitry  54 , the RF communications circuitry  54  uses the second tunable RF filter path  68  and the third tunable RF filter path  110  to provide receive CA. In an alternate embodiment of the RF communications circuitry  54 , tunable RF filters allow for sharing a signal path to provide both an FDD signal path and a TDD signal path, thereby lowering front-end complexity. 
       FIG. 16  shows the RF communications circuitry  54  according to one embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 16  is similar to the RF communications circuitry  54  illustrated in  FIG. 15 , except in the RF communications circuitry  54  illustrated in  FIG. 16 , the third tunable RF filter path  110  is omitted. Additionally, in one embodiment of the RF communications circuitry  54 , the RF receive circuitry  62 , the RF transmit circuitry  64 , and the first RF filter structure  60  are all broadband devices. As such, the RF communications circuitry  54  is broadband circuitry capable of processing RF signals having wide frequency ranges. 
       FIG. 17  shows the RF communications circuitry  54  according to an alternate embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 17  is similar to the RF communications circuitry  54  illustrated in  FIG. 16 , except in the RF communications circuitry  54  illustrated in  FIG. 17 , the RF receive circuitry  62  is omitted and the RF front-end circuitry  58  further includes a first RF front-end circuit  140 , a second RF front-end circuit  142 , and a third RF front-end circuit  144 . 
     The first RF front-end circuit  140  includes the RF transmit circuitry  64 . The second RF front-end circuit  142  includes the first RF filter structure  60 , the first connection node  70 , the second connection node  72 , and the first common connection node  74 . The third RF front-end circuit  144  includes the RF antenna switch  130 . In one embodiment of the first RF front-end circuit  140 , the first RF front-end circuit  140  is a first RF front-end integrated circuit (IC). In one embodiment of the second RF front-end circuit  142 , the second RF front-end circuit  142  is a second RF front-end IC. In one embodiment of the third RF front-end circuit  144 , the third RF front-end circuit  144  is a third RF front-end IC. 
       FIG. 18  shows the RF communications circuitry  54  according to an additional embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 18  is similar to the RF communications circuitry  54  illustrated in  FIG. 16 , except in the RF communications circuitry  54  illustrated in  FIG. 18 , the RF receive circuitry  62  is omitted and the RF front-end circuitry  58  further includes the first RF front-end circuit  140  and the second RF front-end circuit  142 . 
     The first RF front-end circuit  140  includes the RF transmit circuitry  64 . The second RF front-end circuit  142  includes the first RF filter structure  60 , the RF antenna switch  130 , the first connection node  70 , the second connection node  72 , and the first common connection node  74 . In one embodiment of the first RF front-end circuit  140 , the first RF front-end circuit  140  is the first RF front-end IC. In one embodiment of the second RF front-end circuit  142 , the second RF front-end circuit  142  is the second RF front-end IC. 
       FIG. 19  shows the RF communications circuitry  54  according to another embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 19  is similar to the RF communications circuitry  54  illustrated in  FIG. 16 , except in the RF communications circuitry  54  illustrated in  FIG. 19 , the RF receive circuitry  62  is omitted and the RF front-end circuitry  58  further includes the first RF front-end circuit  140 . 
     The first RF front-end circuit  140  includes the RF transmit circuitry  64 , the first RF filter structure  60 , the RF antenna switch  130 , the first connection node  70 , the second connection node  72 , and the first common connection node  74 . In one embodiment of the first RF front-end circuit  140 , the first RF front-end circuit  140  is the first RF front-end IC. 
       FIG. 20  shows the RF communications circuitry  54  according to a further embodiment of the RF communications circuitry  54 . The RF communications circuitry  54  illustrated in  FIG. 20  is a TDD system, which is capable of transmitting and receiving RF signals, but not simultaneously. As such, the RF communications circuitry  54  illustrated in  FIG. 20  is similar to the RF communications circuitry  54  illustrated in  FIG. 4 , except in the RF communications circuitry  54  illustrated in  FIG. 20 , the second tunable RF filter path  68  and the second connection node  72  are omitted, and the RF front-end circuitry  58  further includes an RF transmit/receive switch  146  coupled between the first tunable RF filter path  66  and the RF receive circuitry  62 , and further coupled between the first tunable RF filter path  66  and the RF transmit circuitry  64 . 
     Since the RF communications circuitry  54  does not simultaneously transmit and receive RF signals, the first tunable RF filter path  66  provides front-end transmit filtering when the RF communications circuitry  54  is transmitting RF signals and the first tunable RF filter path  66  provides front-end receive filtering when the RF communications circuitry  54  is receiving RF signals. In this regard, the first tunable RF filter path  66  processes half-duplex signals. 
     The RF transmit/receive switch  146  has a transmit/receive switch common connection node  148 , a transmit/receive switch first connection node  150 , and a transmit/receive switch second connection node  152 . The RF receive circuitry  62  is coupled between the RF system control circuitry  56  and the transmit/receive switch second connection node  152 . The RF transmit circuitry  64  is coupled between the RF system control circuitry  56  and the transmit/receive switch first connection node  150 . The first connection node  70  is coupled to the transmit/receive switch common connection node  148 . 
     The RF system control circuitry  56  provides a switch control signal SCS to the RF transmit/receive switch  146 . As such, the RF system control circuitry  56  selects either the transmit/receive switch first connection node  150  or the transmit/receive switch second connection node  152  to be coupled to the transmit/receive switch common connection node  148  using the switch control signal SCS. Therefore, when the RF communications circuitry  54  is transmitting RF signals, the RF transmit circuitry  64  is coupled to the first tunable RF filter path  66  and the RF receive circuitry  62  is not coupled to the first tunable RF filter path  66 . Conversely, when the RF communications circuitry  54  is receiving RF signals, the RF receive circuitry  62  is coupled to the first tunable RF filter path  66  and the RF transmit circuitry  64  is not coupled to the first tunable RF filter path  66 . 
       FIG. 21  illustrates an exemplary embodiment of the first RF filter structure  60 . The first RF filter structure  60  includes a plurality of resonators (referred to generically as elements R and specifically as elements R(i,j), where an integer i indicates a row position and an integer j indicates a column position, where 1≦i≦M, 1≦j≦N and M is any integer greater than 1 and N is any integer greater than to 1. It should be noted that in alternative embodiments the number of resonators R in each row and column may be the same or different). The first tunable RF filter path  66  includes row  1  of weakly coupled resonators R( 1 , 1 ), R( 1 , 2 ) through (R( 1 ,N). All of the weakly coupled resonators R( 1 , 1 ), R( 1 , 2 ) through (R( 1 ,N) are weakly coupled to one another. Furthermore, the first tunable RF filter path  66  is electrically connected between terminal  200  and terminal  202 . In this manner, the first tunable RF filter path  66  is configured to receive RF signals and output filtered RF signals. The second tunable RF filter path  68  includes row M of weakly coupled resonators R(M, 1 ), R(M, 2 ) through R(M,N). All of the weakly coupled resonators R(M, 1 ), R(M, 2 ) through R(M,N) are weakly coupled to one another. Furthermore, the second tunable RF filter path  68  is electrically connected between terminal  204  and terminal  206 . In this manner, the second tunable RF filter path  68  is configured to receive RF signals and output filtered RF signals. It should be noted that the first RF filter structure  60  may include any number of tunable RF filter paths, such as, for example, the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124 , described above with respect to  FIGS. 11-14 . Each of the resonators R may be a tunable resonator, which allows for a resonant frequency of each of the resonators R to be varied to along a frequency range. In some embodiments, not all of the couplings between the resonators R are weak. A hybrid architecture having at least one pair of weakly coupled resonators R and strongly or moderately coupled resonators R is also possible. 
     Cross-coupling capacitive structures C are electrically connected to and between the resonators R. In this embodiment, each of the cross-coupling capacitive structures C is a variable cross-coupling capacitive structure, such as a varactor or an array of capacitors. To be independent, the magnetic couplings may be negligible. Alternatively, the cross-coupling capacitive structures C may simply be provided by a capacitor with a fixed capacitance. With regard to the exemplary embodiment shown in  FIG. 21 , the tunable RF filter paths of the first RF filter structure  60  are independent of one another. As such, the first tunable RF filter path  66  and the second tunable RF filter path  68  are independent of one another and thus do not have cross-coupling capacitive structures C between their resonators. Thus, in this embodiment, the cross-coupling capacitive structures C do not connect any of the weakly coupled resonators R( 1 , 1 ), R( 1 , 2 ) through (R( 1 ,N) to any of the weakly coupled resonators R(M, 1 ), R(M, 2 ) through (R(M,N). This provides increased isolation between the first tunable RF filter path  66  and the second tunable RF filter path  68 . In general, energy transfer between two weakly coupled resonators R in the first tunable RF filter path  66  and the second tunable RF filter path  68  may be provided by multiple energy transfer components. For example, energy may be transferred between the resonators R only through mutual magnetic coupling, only through mutual electric coupling, or through both mutual electric coupling and mutual magnetic coupling. Ideally, all of the mutual coupling coefficients are provided as designed, but in practice, the mutual coupling coefficients also be the result of parasitics. The inductors of the resonators R may also have magnetic coupling between them. A total coupling between the resonators R is given by the sum of magnetic and electric coupling. 
     In order to provide the transfer functions of the tunable RF filter paths  66 ,  68  with high out-of-band attenuation and a relatively low filter order, the tunable RF filter paths  66 ,  68  are configured to adjust notches in the transfer function, which are provided by the resonators R within the tunable RF filter paths  66 ,  68 . The notches can be provided using parallel tanks connected in series or in shunt along a signal path of the first tunable RF filter path  66 . To provide the notches, the parallel tanks operate approximately as an open circuit or as short circuits at certain frequencies. The notches can also be provided using multi-signal path cancellation. In this case, the tunable RF filter paths  66 ,  68  may be smaller and/or have fewer inductors. To tune the total mutual coupling coefficients between the resonators R towards a desired value, the tunable RF filter paths  66 ,  68  are configured to vary variable electric coupling coefficients so that parasitic couplings between the resonators R in the tunable RF filter paths  66 ,  68  are absorbed into a desired frequency transfer function. 
       FIG. 22  illustrates an exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 22  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 22 . The first tunable RF filter path  66  shown in  FIG. 22  includes an embodiment of the resonator R( 1 , 1 ) and an embodiment of the resonator R( 1 , 2 ). The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled to one another. More specifically, the resonator R( 1 , 1 ) includes an inductor  208  and a capacitive structure  210 . The resonator R( 1 , 2 ) includes an inductor  212 , a capacitive structure  214 , and a capacitive structure  216 . 
     The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are a pair of weakly coupled resonators. The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled by providing the inductor  208  and the inductor  212  such that the inductor  208  and the inductor  212  are weakly magnetically coupled. Although the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled, the inductor  212  has a maximum lateral width and a displacement between the inductor  208  and the inductor  212  is less than or equal to half the maximum lateral width of the inductor  212 . As such, the inductor  208  and the inductor  212  are relatively close to one another. The displacement between the inductor  208  and the inductor  212  may be measured from a geometric centroid of the inductor  208  to a geometric centroid of the inductor  212 . The maximum lateral width may be a maximum dimension of the inductor  212  along a plane defined by its largest winding. The weak coupling between the inductor  208  and the inductor  212  is obtained through topological techniques. For example, the inductor  208  and the inductor  212  may be fully or partially aligned, where winding(s) of the inductor  208  and winding(s) of the inductor  212  are configured to provide weak coupling through cancellation. Alternatively or additionally, a plane defining an orientation of the winding(s) of the inductor  208  and a plane defining an orientation of the winding(s) of the inductor  212  may be fully or partially orthogonal to one another. Some of the magnetic couplings between the resonators R can be unidirectional (passive or active). This can significantly improve isolation (e.g., transmit and receive isolation in duplexers). 
     To maximize the quality (Q) factor of the tunable RF filter paths  66  through  68 , most of the total mutual coupling should be realized magnetically, and only fine-tuning is provided electrically. This also helps to reduce common-mode signal transfer in the differential resonators and thus keeps the Q factor high. While the magnetic coupling can be adjusted only statically, with a new layout design, the electric coupling can be tuned on the fly (after fabrication). The filter characteristics (e.g., bias network structure, resonator capacitance) can be adjusted based on given coupling coefficients to maximize filter performance. 
     To provide a tuning range to tune a transfer function of the first tunable RF filter path  66  and provide a fast roll-off from a low-frequency side to a high-frequency side of the transfer function, the first tunable RF filter path  66  is configured to change a sign of a total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). Accordingly, the first tunable RF filter path  66  includes a cross-coupling capacitive structure C(P 1 ) and a cross-coupling capacitive structure C(N 1 ). The cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure C(N 1 ) are embodiments of the cross-coupling capacitive structures C described above with regard to  FIG. 21 . As shown in  FIG. 22 , the cross-coupling capacitive structure C(P 1 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide a positive coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The cross-coupling capacitive structure C(P 1 ) is a variable cross-coupling capacitive structure configured to vary the positive coupling coefficient provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The cross-coupling capacitive structure C(N 1 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide a negative coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The cross-coupling capacitive structure C(N 1 ) is a variable cross-coupling capacitive structure configured to vary the negative coupling coefficient provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The arrangement of the cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure C(N 1 ) shown in  FIG. 22  is a V-bridge structure. In alternative embodiments, some or all of the cross-coupling capacitive structures is fixed (not variable). 
     In the resonator R( 1 , 1 ), the inductor  208  and the capacitive structure  210  are electrically connected in parallel. More specifically, the inductor  208  has an end  217  and an end  218 , which are disposed opposite to one another. The ends  217 ,  218  are each electrically connected to the capacitive structure  210 , which is grounded. Thus, the resonator R( 1 , 1 ) is a single-ended resonator. On the other hand, the inductor  212  is electrically connected between the capacitive structure  214  and the capacitive structure  216 . More specifically, the inductor  212  has an end  220  and an end  222 , which are disposed opposite to one another. The end  220  is electrically connected to the capacitive structure  214  and the end  222  is electrically connected to the capacitive structure  216 . Both the capacitive structure  214  and the capacitive structure  216  are grounded. Thus, the resonator R( 1 , 2 ) is a differential resonator. In an alternative, an inductor with a center tap can be used. The tap can be connected to ground and only a single capacitive structure can be used. In yet another embodiment, both an inductor and a capacitive structure may have a center tap that is grounded. In still another embodiment, neither the inductor nor the capacitive structure may have a grounded center tap. 
     The inductor  208  is magnetically coupled to the inductor  212  such that an RF signal received at the end  217  of the inductor  208  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal being transmitted out the end  220  of the inductor  212  with the same voltage polarity. Also, the inductor  212  is magnetically coupled to the inductor  208  such that an RF signal received at the end  220  of the inductor  212  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal being transmitted out the end  217  of the inductor  208  with the same voltage polarity. This is indicated in  FIG. 22  by the dot convention where a dot is placed at the end  217  of the inductor  208  and a dot is placed at the end  220  of the inductor  212 . By using two independent and adjustable coupling coefficients (i.e., the positive coupling coefficient and the negative coupling coefficient) with the resonator R( 1 , 2 ) (i.e., the differential resonator), the transfer function of the first tunable RF filter path  66  is provided so as to be fully adjustable. More specifically, the inductors  208 ,  212  may be magnetically coupled so as to have a low magnetic coupling coefficient through field cancellation, with the variable positive coupling coefficient and the variable negative coupling coefficient. In this case, the inductor  208  and the inductor  212  are arranged such that a mutual magnetic coupling between the inductor  208  and the inductor  212  cancel. Alternatively, the inductor  208  and the inductor  212  are arranged such that the inductor  212  reduces a mutual magnetic coupling coefficient of the inductor  208 . With respect to the magnetic coupling coefficient, the variable positive coupling coefficient is a variable positive electric coupling coefficient and the variable negative coupling coefficient is a variable negative electric coupling coefficient. The variable positive electric coupling coefficient and the variable negative electric coupling coefficient oppose each other to create a tunable filter characteristic. 
     The resonator R( 1 , 2 ) is operably associated with the resonator R( 1 , 1 ) such that an energy transfer factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is less than 10%. A total mutual coupling between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is provided by a sum total of the mutual magnetic factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) and the mutual electric coupling coefficients between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). In this embodiment, the mutual magnetic coupling coefficient between the inductor  208  and the inductor  212  is a fixed mutual magnetic coupling coefficient. Although embodiments of the resonators R( 1 , 1 ), R( 1 , 2 ) may be provided so as to provide a variable magnetic coupling coefficient between the resonators R( 1 , 1 ), R( 1 , 2 ), embodiments of the resonators R( 1 , 1 ), R( 1 , 2 ) that provide variable magnetic couplings can be costly and difficult to realize. However, providing variable electric coupling coefficients (i.e., the variable positive electric coupling coefficient and the variable electric negative coupling coefficient) is easier and more economical. Thus, using the cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure C(N 1 ) to provide the variable positive electric coupling coefficient and the variable electric negative coupling coefficient is an economical technique for providing a tunable filter characteristic between the resonators R( 1 , 1 ), R( 1 , 2 ). Furthermore, since the mutual magnetic coupling coefficient between the inductor  208  and the inductor  212  is fixed, the first tunable RF filter path  66  has lower insertion losses. 
     In the embodiment shown in  FIG. 22 , the inductor  208  and the  212  inductor are the same size. Alternatively, the inductor  208  and the inductor  212  may be different sizes. For example, the inductor  212  may be smaller than the inductor  208 . By determining a distance between the inductor  208  and the inductor  212 , the magnetic coupling coefficient between the inductor  208  and the inductor  212  can be set. With regard to the inductors  208 ,  212  shown in  FIG. 22 , the inductor  208  may be a folded inductor configured to generate a first confined magnetic field, while the inductor  212  may be a folded inductor configured to generate a second confined magnetic field. Magnetic field lines of the first confined magnetic field and of the second confined magnetic field that are external to the inductor  208  and inductor  212  are cancelled by opposing magnetic field lines in all directions. When the inductor  208  and the inductor  212  are folded inductors, the folded inductors can be stacked. This allows building the first tunable RF filter path  66  such that several inductors  208 ,  212  are stacked. Furthermore, this arrangement allows for a specially sized interconnect structure that electrically connects the inductors  208 ,  212  to the capacitive structure  210 , the capacitive structure  214 , the capacitive structure  216 , the cross-coupling capacitive structure C(P 1 ), and the cross-coupling capacitive structure C(N 1 ). The specially sized interconnect increases the Q factor of the capacitive structure  210 , the capacitive structure  214 , the capacitive structure  216 , the cross-coupling capacitive structure C(P 1 ), and the cross-coupling capacitive structure C(N 1 ), and allows for precise control of their variable capacitances. Weakly coupled filters can also be realized with planar field cancellation structures. 
       FIG. 23  illustrates an exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 23  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 23 . The first tunable RF filter path  66  shown in  FIG. 23  includes an embodiment of the resonator R( 1 , 1 ) and an embodiment of the resonator R( 1 , 2 ). The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled to one another. The embodiment of the resonator R( 1 , 2 ) is the same as the embodiment of the resonator R( 1 , 2 ) shown in  FIG. 22 . Thus, the resonator R( 1 , 2 ) shown in  FIG. 23  is a differential resonator that includes the inductor  212 , the capacitive structure  214 , and the capacitive structure  216 . Additionally, like the embodiment of the resonator R( 1 , 1 ) shown in  FIG. 22 , the embodiment of the resonator R( 1 , 1 ) shown in  FIG. 23  includes the inductor  208  and the capacitive structure  210 . However, in this embodiment, the resonator R( 1 , 1 ) shown in  FIG. 23  is a differential resonator and further includes a capacitive structure  224 . More specifically, the end  217  of the inductor  208  is electrically connected to the capacitive structure  210  and the end  218  of the inductor  208  is electrically connected to the capacitive structure  224 . Both the capacitive structure  210  and the capacitive structure  224  are grounded. Like the capacitive structure  210 , the capacitive structure  224  is also a variable capacitive structure, such as a programmable array of capacitors or a varactor. Alternatively, a center tap of an inductor may be grounded. In yet another embodiment, the inductor and a capacitive structure may be RF floating (a low-resistance connection to ground). 
     The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are a pair of weakly coupled resonators. Like the first tunable RF filter path  66  shown in  FIG. 22 , the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled by providing the inductor  208  and the inductor  212  such that the inductor  208  and the inductor  212  are weakly coupled. Thus, the inductor  208  and the inductor  212  may have a magnetic coupling coefficient that is less than or equal to approximately 0.3. Although the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled, a displacement between the inductor  208  and the inductor  212  is less than or equal to half the maximum lateral width of the inductor  212 . As such, the inductor  208  and the inductor  212  are relatively close to one another. The displacement between the inductor  208  and the inductor  212  may be measured from a geometric centroid of the inductor  208  to a geometric centroid of the inductor  212 . The maximum lateral width may be a maximum dimension of the inductor  212  along a plane defined by its largest winding. 
     The weak coupling between the inductor  208  and the inductor  212  is obtained through topological techniques. For example, the inductor  208  and the inductor  212  may be fully or partially aligned, where winding(s) of the inductor  208  and winding(s) of the inductor  212  are configured to provide weak coupling through cancellation. Alternatively or additionally, a plane defining an orientation of the windings of the inductor  208  and a plane defining an orientation of the windings of the inductor  212  may be fully or partially orthogonal to one another. 
     The resonator R( 1 , 2 ) is operably associated with the resonator R( 1 , 1 ) such that an energy transfer factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is less than 10%. To provide a tuning range to tune a transfer function of the first tunable RF filter path  66  such to provide a fast roll-off from a low-frequency side to a high-frequency side requires changing a sign of the total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). Like the embodiment of the first tunable RF filter path  66  shown in  FIG. 22 , the first tunable RF filter path  66  shown in  FIG. 23  includes the cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure C(N 1 ). The cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure C(N 1 ) are arranged in the same manner described above with respect to  FIG. 22 . However, in this embodiment, the first tunable RF filter path  66  shown in  FIG. 23  also includes a cross-coupling capacitive structure C(P 2 ) and a cross-coupling capacitive structure C(N 2 ). The cross-coupling capacitive structure C(P 2 ) and the cross-coupling capacitive structure C(N 2 ) are also embodiments of the cross-coupling capacitive structures C described above with regard to  FIG. 21 . 
     As described above with respect to  FIG. 22 , the cross-coupling capacitive structure C(P 1 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide the positive coupling coefficient (i.e., the variable positive electric coupling coefficient) between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). Also as described above with respect to  FIG. 22 , the cross-coupling capacitive structure C(N 1 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide the negative coupling coefficient (i.e., the variable negative electric coupling coefficient) between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). With regard to the cross-coupling capacitive structure C(P 2 ), the cross-coupling capacitive structure C(P 2 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide another positive coupling coefficient (i.e., another variable positive electric coupling coefficient) between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). In this embodiment, the cross-coupling capacitive structure C(P 2 ) is electrically connected between the end  218  of the inductor  208  and the end  222  of the inductor  212 . The cross-coupling capacitive structure C(P 2 ) is a variable cross-coupling capacitive structure configured to vary the other positive coupling coefficient (i.e., the other variable positive electric coupling coefficient) provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). With regard to the cross-coupling capacitive structure C(N 2 ), the cross-coupling capacitive structure C(N 2 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) so as to provide another negative coupling coefficient (i.e., another variable negative electric coupling coefficient) between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). In this embodiment, the cross-coupling capacitive structure C(N 2 ) is electrically connected between the end  218  of the inductor  208  and the end  220  of the inductor  212 . The cross-coupling capacitive structure C(N 2 ) is a variable cross-coupling capacitive structure configured to vary the negative coupling coefficient (i.e., the other variable negative electric coupling coefficient) provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The arrangement of the cross-coupling capacitive structure C(P 1 ), the cross-coupling capacitive structure C(N 1 ), the cross-coupling capacitive structure C(P 2 ), and the cross-coupling capacitive structure C(N 2 ) shown in  FIG. 23  is an X-bridge structure. 
     As shown in  FIG. 23 , the resonator R( 1 , 2 ) is operably associated with the resonator R( 1 , 1 ) such that an energy transfer factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is less than 10%. The total mutual coupling between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is provided by a sum total of the mutual magnetic factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) and the mutual electric coupling coefficients between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). Thus, in this embodiment, the total mutual coupling between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is provided by the sum total of the mutual magnetic coupling coefficient, the variable positive electric coupling coefficient provided by the cross-coupling capacitive structure C(P 1 ), the variable negative electric coupling coefficient provided by the cross-coupling capacitive structure C(N 1 ), the other variable positive electric coupling coefficient provided by the cross-coupling capacitive structure C(P 2 ), and the other variable negative electric coupling coefficient provided by the cross-coupling capacitive structure C(N 2 ). 
       FIG. 24  illustrates an exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 24  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 24 . The first tunable RF filter path  66  shown in  FIG. 24  includes an embodiment of the resonator R( 1 , 1 ) and an embodiment of the resonator R( 1 , 2 ). The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled to one another. The embodiment of the resonator R( 1 , 1 ) is the same as the embodiment of the resonator R( 1 , 1 ) shown in  FIG. 22 . Thus, the resonator R( 1 , 1 ) shown in  FIG. 24  is a single-ended resonator that includes the inductor  208  and the capacitive structure  210 . Additionally, like the embodiment of the resonator R( 1 , 2 ) shown in  FIG. 22 , the embodiment of the resonator R( 1 , 2 ) shown in  FIG. 24  includes the inductor  212  and the capacitive structure  214 . However, in this embodiment, the resonator R( 1 , 2 ) shown in  FIG. 24  is a single-ended resonator. More specifically, the end  220  and the end  222  of the inductor  212  are each electrically connected to the capacitive structure  214 , which is grounded. 
     The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are a pair of weakly coupled resonators. Like the first tunable RF filter path  66  shown in  FIG. 22 , the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled by providing the inductor  208  and the inductor  212  such that the inductor  208  and the inductor  212  are weakly coupled. Thus, the inductor  208  and the inductor  212  may have a magnetic coupling coefficient that is less than or equal to approximately 0.3. Although the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled, the displacement between the inductor  208  and the inductor  212  is less than or equal to half the maximum lateral width of the inductor  212 . As such, the inductor  208  and the inductor  212  are relatively close to one another. The displacement between the inductor  208  and the inductor  212  may be measured from the geometric centroid of the inductor  208  to the geometric centroid of the inductor  212 . The maximum lateral width may be a maximum dimension of the inductor  212  along a plane defined by its largest winding. The weak coupling between the inductor  208  and the inductor  212  is obtained through topological techniques. For example, the inductor  208  and the inductor  212  may be fully or partially aligned, where winding(s) of the inductor  208  and winding(s) of the inductor  212  are configured to provide weak coupling through cancellation. Alternatively or additionally, a plane defining an orientation of the windings of the inductor  208  and a plane defining an orientation of the windings of the inductor  212  may be fully or partially orthogonal to one another. 
     The resonator R( 1 , 2 ) is operably associated with the resonator R( 1 , 1 ) such that an energy transfer factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is less than 10%. To provide a tuning range to tune a transfer function of the first tunable RF filter path  66  and provide a fast roll-off from a low-frequency side to a high-frequency side of the transfer function, the first tunable RF filter path  66  is configured to change a sign of a total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). However, in this embodiment, the first tunable RF filter path  66  shown in  FIG. 24  only includes the cross-coupling capacitive structure C(P 1 ), which is electrically connected between the end  217  of the inductor  208  and the end  220  of the inductor  212 . As discussed above with respect to  FIGS. 22 and 23 , the cross-coupling capacitive structure C(P 1 ) is a variable cross-coupling capacitive structure configured to vary the positive coupling coefficient (i.e., the variable positive electric coupling coefficient) provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). Thus, in order to allow for the sign of the total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) to be changed, the inductor  208  and the inductor  212  are arranged so as to provide a fixed negative mutual magnetic coupling coefficient between the inductor  208  of the resonator R( 1 , 1 ) and the inductor  212  of the resonator R( 1 , 2 ). As such, varying the variable positive electric coupling coefficient allows for the sign of the total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) to be changed using only the cross-coupling capacitive structure C(P 1 ). 
     As such, in this embodiment, the inductor  208  is magnetically coupled to the inductor  212  such that an RF signal received at the end  217  of the inductor  208  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal with the same voltage polarity being transmitted out the end  222  of the inductor  212 . In addition, the inductor  212  is magnetically coupled to the inductor  208  such that an RF signal received at the end  222  of the inductor  212  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal with the same voltage polarity being transmitted out the end  217  of the inductor  208 . This is indicated in  FIG. 24  by the dot convention where a dot is placed at the end  217  of the inductor  208  and a dot is placed at the end  222  of the inductor  212 . By using the fixed negative mutual magnetic coupling coefficient and the variable positive electric coupling coefficient, the transfer function of the first tunable RF filter path  66  is provided so to be fully adjustable. The arrangement of the cross-coupling capacitive structure C(P 1 ) shown in  FIG. 24  is a single positive bridge structure. 
       FIG. 25  illustrates another exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 25  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 25 . The first tunable RF filter path  66  shown in  FIG. 25  includes an embodiment of the resonator R( 1 , 1 ) and an embodiment of the resonator R( 1 , 2 ). The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled to one another. The embodiment of the resonator R( 1 , 1 ) is the same as the embodiment of the resonator R( 1 , 1 ) shown in  FIG. 22 . Thus, the resonator R( 1 , 1 ) shown in  FIG. 25  is a single-ended resonator that includes the inductor  208  and the capacitive structure  210 , which are arranged in the same manner described above with respect to  FIG. 22 . Like the resonator R( 1 , 2 ) shown in  FIG. 24 , the resonator R( 1 , 2 ) shown in  FIG. 25  is a single-ended resonator that includes the inductor  212  and the capacitive structure  214 . However, the inductor  208  shown in  FIG. 25  is magnetically coupled to the inductor  212  such that an RF signal received at the end  217  of the inductor  208  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal with the same voltage polarity being transmitted out the end  220  of the inductor  212 . Also, the inductor  212  shown in  FIG. 25  is magnetically coupled to the inductor  208  such that an RF signal received at the end  220  of the inductor  212  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in a filtered RF signal with the same voltage polarity being transmitted out the end  217  of the inductor  208 . This is indicated in  FIG. 25  by the dot convention where a dot is placed at the end  217  of the inductor  208  and a dot is placed at the end  220  of the inductor  212 . In alternative embodiments, the resonator R( 1 , 2 ) is a differential resonator. In yet another alternative embodiment, the resonator R( 1 , 1 ) is a single-ended resonator while the resonator R( 1 , 2 ) is a differential resonator. 
     The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are a pair of weakly coupled resonators. Like the first tunable RF filter path  66  shown in  FIG. 22 , the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled by providing the inductor  208  and the inductor  212  such that the inductor  208  and the inductor  212  are weakly coupled. Thus, the inductor  208  and the inductor  212  may have a fixed magnetic coupling coefficient that is less than or equal to approximately 0.3. Although the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are weakly coupled, a displacement between the inductor  208  and the inductor  212  is less than or equal to half the maximum lateral width of the inductor  212 . As such, the inductor  208  and the inductor  212  are relatively close to one another. The displacement between the inductor  208  and the inductor  212  may be measured from a geometric centroid of the inductor  208  to a geometric centroid of the inductor  212 . The maximum lateral width may be a maximum dimension of the inductor  212  along a plane defined by its largest winding. 
     The weak coupling between the inductor  208  and the inductor  212  is obtained through topological techniques. For example, the inductor  208  and the inductor  212  may be fully or partially aligned, where winding(s) of the inductor  208  and winding(s) of the inductor  212  are configured to provide weak coupling through cancellation. Alternatively or additionally, a plane defining an orientation of the windings of the inductor  208  and a plane defining an orientation of the windings of the inductor  212  may be fully or partially orthogonal to one another. 
     The resonator R( 1 , 2 ) is operably associated with the resonator R( 1 , 1 ) such that an energy transfer factor between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) is less than 10%. To provide a tuning range to tune the transfer function of the first tunable RF filter path  66  and to provide a fast roll-off from the low-frequency side to the high-frequency side of the transfer function, the first tunable RF filter path  66  is configured to change the sign of the total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). In this embodiment, the first tunable RF filter path  66  shown in  FIG. 25  includes a cross-coupling capacitive structure C(PH 1 ), a cross-coupling capacitive structure (CNH 1 ), a cross-coupling capacitive structure C(I 1 ), a cross-coupling capacitive structure C(PH 2 ), and a cross-coupling capacitive structure C(NH 2 ). The cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure (CNH 1 ), the cross-coupling capacitive structure C(I 1 ), the cross-coupling capacitive structure C(PH 2 ), and the cross-coupling capacitive structure C(NH 2 ) are also embodiments of the cross-coupling capacitive structures C described above with regard to  FIG. 21 . 
     The cross-coupling capacitive structure C(PH 1 ) and the cross-coupling capacitive structure C(NH 1 ) are arranged to form a first capacitive voltage divider. The first capacitive voltage divider is electrically connected to the resonator R( 1 , 1 ). More specifically, the cross-coupling capacitive structure C(PH 1 ) is electrically connected between the end  217  of the inductor  208  and a common connection node H 1 . The cross-coupling capacitive structure C(NH 1 ) is electrically connected between the end  218  of the inductor  208  and the common connection node H 1 . Additionally, the cross-coupling capacitive structure C(PH 2 ) and the cross-coupling capacitive structure C(NH 2 ) are arranged to form a second capacitive voltage divider. The second capacitive voltage divider is electrically connected to the resonator R( 1 , 2 ). More specifically, the cross-coupling capacitive structure C(PH 2 ) is electrically connected between the end  220  of the inductor  212  and a common connection node H 2 . The cross-coupling capacitive structure C(NH 2 ) is electrically connected between the end  222  of the inductor  212  and the common connection node H 2 . As shown in  FIG. 25 , the cross-coupling capacitive structure C(I 1 ) is electrically connected between the first capacitive voltage divider and the second capacitive voltage divider. More specifically, the cross-coupling capacitive structure C(I 1 ) is electrically connected between the common connection node H 1  and the common connection node H 2 . The arrangement of the cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure C(NH 1 ), the cross-coupling capacitive structure C(PH 2 ), the cross-coupling capacitive structure C(NH 2 ), and the cross-coupling capacitive structure C(I 1 ) shown in  FIG. 25  is an H-bridge structure. In an alternative H-bridge structure, the cross-coupling capacitive structure C(I 1 ) is not provided and instead there is a short between the common connection node H 1  and the common connection node H 2 . In addition, a center tap of the inductor  208  may be grounded and/or the common connection node H 1  may be grounded. Finally, a high impedance to ground may be provided at the common connection node H 1 . 
     With regard to the first capacitive voltage divider, the cross-coupling capacitive structure C(PH 1 ) is a variable cross-coupling capacitive structure configured to vary a first variable positive electric coupling coefficient provided between the resonator R( 1 , 1 ) and the common connection node H 1 . The cross-coupling capacitive structure C(NH 1 ) is a variable cross-coupling capacitive structure configured to vary a first variable negative electric coupling coefficient provided between the resonator R( 1 , 1 ) and the common connection node H 1 . Thus, a mutual electric coupling coefficient of the resonator R( 1 , 1 ) is approximately equal to the first variable positive electric coupling coefficient and the first variable negative electric coupling coefficient. 
     With regard to the second capacitive voltage divider, the cross-coupling capacitive structure C(PH 2 ) is a variable cross-coupling capacitive structure configured to vary a second variable positive electric coupling coefficient provided between the resonator R( 1 , 2 ) and the common connection node H 2 . The cross-coupling capacitive structure C(NH 2 ) is a variable cross-coupling capacitive structure configured to vary a second variable negative electric coupling coefficient provided between the resonator R( 1 , 2 ) and the common connection node H 2 . Thus, a mutual electric coupling coefficient of the resonator R( 1 , 2 ) is approximately equal to the second variable positive electric coupling coefficient and the second variable negative electric coupling coefficient. Furthermore, the cross-coupling capacitive structure C(I 1 ) is a variable cross-coupling capacitive structure configured to vary a first variable intermediate electric coupling coefficient provided between the common connection node H 1  and the common connection node H 2 . The first tunable RF filter path  66  shown in  FIG. 25  thus has a total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) equal to the sum total of the mutual magnetic coupling coefficient between the inductor  208  and the inductor  212 , the mutual electric coupling coefficient of the resonator R( 1 , 1 ), the mutual electric coupling coefficient of the resonator R( 1 , 2 ), and the first variable intermediate electric coupling coefficient provided between the common connection node H 1  and the common connection node H 2 . In alternative embodiments, cross-coupling capacitive structures with fixed capacitances are provided. 
     In one embodiment, the cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure C(NH 1 ), the cross-coupling capacitive structure C(PH 2 ), the cross-coupling capacitive structure C(NH 2 ), and the cross-coupling capacitive structure C(I 1 ) may each be provided as a varactor. However, the cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure C(NH 1 ), the cross-coupling capacitive structure C(PH 2 ), the cross-coupling capacitive structure C(NH 2 ), and the cross-coupling capacitive structure C(I 1 ) may each be provided as a programmable array of capacitors in order to reduce insertion losses and improve linearity. The cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure C(NH 1 ), the cross-coupling capacitive structure C(PH 2 ), the cross-coupling capacitive structure C(NH 2 ), and the cross-coupling capacitive structure C(I 1 ) can also be any combination of suitable variable cross-coupling capacitive structures, such as combinations of varactors and programmable arrays of capacitors. Although the H-bridge structure can provide good linearity and low insertion losses, the H-bridge structure can also suffer from common-mode signal transfer. 
       FIG. 26  illustrates yet another exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 26  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 26 . The first tunable RF filter path  66  shown in  FIG. 26  can be used to ameliorate the common-mode signal transfer of the H-bridge structure shown in  FIG. 25 . More specifically, the first tunable RF filter path  66  shown in  FIG. 26  includes the same embodiment of the resonator R( 1 , 1 ) and the same embodiment of the resonator R( 1 , 2 ) described above with respect to  FIG. 25 . Furthermore, the first tunable RF filter path  66  shown in  FIG. 26  includes the first capacitive voltage divider with the cross-coupling capacitive structure C(PH 1 ) and the cross-coupling capacitive structure C(NH 1 ) described above with respect to  FIG. 25 , the second capacitive voltage divider with the cross-coupling capacitive structure C(PH 2 ) and the cross-coupling capacitive structure (CNH 2 ) described above with respect to  FIG. 25 , and the cross-coupling capacitive structure C(I 1 ) described above with respect to  FIG. 25 . However, in this embodiment, the first tunable RF filter path  66  shown in  FIG. 26  also includes a cross-coupling capacitive structure C(PH 3 ), a cross-coupling capacitive structure (CNH 3 ), a cross-coupling capacitive structure C(I 2 ), a cross-coupling capacitive structure C(PH 4 ), and a cross-coupling capacitive structure C(NH 4 ). The cross-coupling capacitive structure C(PH 3 ), the cross-coupling capacitive structure (CNH 3 ), the cross-coupling capacitive structure C(I 2 ), the cross-coupling capacitive structure C(PH 4 ), and the cross-coupling capacitive structure C(NH 4 ) are also embodiments of the cross-coupling capacitive structures C described above with regard to  FIG. 21 . 
     As shown in  FIG. 26 , the cross-coupling capacitive structure C(PH 3 ) and the cross-coupling capacitive structure C(NH 3 ) are arranged to form a third capacitive voltage divider. The third capacitive voltage divider is electrically connected to the resonator R( 1 , 1 ). More specifically, the cross-coupling capacitive structure C(PH 3 ) is electrically connected between the end  217  of the inductor  208  and a common connection node H 3 . The cross-coupling capacitive structure C(NH 3 ) is electrically connected between the end  218  of the inductor  208  and the common connection node H 3 . Additionally, the cross-coupling capacitive structure C(PH 4 ) and the cross-coupling capacitive structure C(NH 4 ) are arranged to form a fourth capacitive voltage divider. The fourth capacitive voltage divider is electrically connected to the resonator R( 1 , 2 ). More specifically, the cross-coupling capacitive structure C(PH 4 ) is electrically connected between the end  220  of the inductor  212  and a common connection node H 4 . The cross-coupling capacitive structure C(NH 4 ) is electrically connected between the end  222  of the inductor  212  and the common connection node H 4 . As shown in  FIG. 26 , the cross-coupling capacitive structure C(I 2 ) is electrically connected between first capacitive voltage divider and the second capacitive voltage divider. More specifically, the cross-coupling capacitive structure C(I 2 ) is electrically connected between the common connection node H 3  and the common connection node H 4 . Alternatively, the cross-coupling capacitive structure C(I 1 ) and the cross-coupling capacitive structure C(I 2 ) can be replaced with shorts. The arrangement of the cross-coupling capacitive structure C(PH 1 ), the cross-coupling capacitive structure C(NH 1 ), the cross-coupling capacitive structure C(PH 2 ), the cross-coupling capacitive structure C(NH 2 ), the cross-coupling capacitive structure C(I 1 ), the cross-coupling capacitive structure C(PH 3 ), the cross-coupling capacitive structure C(NH 3 ), the cross-coupling capacitive structure C(PH 4 ), the cross-coupling capacitive structure C(NH 4 ), and the cross-coupling capacitive structure C(I 2 ) shown in  FIG. 26  is a double H-bridge structure. 
     With regard to the third capacitive voltage divider, the cross-coupling capacitive structure C(PH 3 ) is a variable cross-coupling capacitive structure configured to vary a third variable positive electric coupling coefficient provided between the resonator R( 1 , 1 ) and the common connection node H 3 . The cross-coupling capacitive structure C(NH 3 ) is a variable cross-coupling capacitive structure configured to vary a third variable negative electric coupling coefficient provided between the resonator R( 1 , 1 ) and the common connection node H 3 . Thus, a mutual electric coupling coefficient of the resonator R( 1 , 1 ) is approximately equal to the first variable positive electric coupling coefficient, the third variable positive electric coupling coefficient, the first variable negative electric coupling coefficient and the third variable negative electric coupling coefficient. 
     With regard to the fourth capacitive voltage divider, the cross-coupling capacitive structure C(PH 4 ) is a variable cross-coupling capacitive structure configured to vary a fourth variable positive electric coupling coefficient provided between the resonator R( 1 , 2 ) and the common connection node H 4 . The cross-coupling capacitive structure C(NH 4 ) is a variable cross-coupling capacitive structure configured to vary a fourth variable negative electric coupling coefficient provided between the resonator R( 1 , 2 ) and the common connection node H 4 . Thus, a mutual electric coupling coefficient of the resonator R( 1 , 2 ) is approximately equal to the second variable positive electric coupling coefficient, the fourth variable positive coupling coefficient, the second variable negative coupling coefficient, and the fourth variable negative electric coupling coefficient. Furthermore, the cross-coupling capacitive structure C(I 2 ) is a variable cross-coupling capacitive structure configured to vary a second variable intermediate electric coupling coefficient provided between the common connection node H 3  and the common connection node H 4 . The first tunable RF filter path  66  shown in  FIG. 26  thus has a total mutual coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) equal to the sum total of the mutual magnetic coupling coefficient between the inductor  208  and the inductor  212 , the mutual electric coupling coefficient of the resonator R( 1 , 1 ), the mutual electric coupling coefficient of the resonator R( 1 , 2 ), the first variable intermediate electric coupling coefficient provided between the common connection node H 1  and the common connection node H 2  and the second variable intermediate electric coupling coefficient provided between the common connection node H 3  and the common connection node H 4 . The double H-bridge structure thus includes two H-bridge structures. The two H-bridge structures allow for common-mode signal transfers of the two H-bridge structures to oppose one another and thereby be reduced and even cancelled. 
       FIG. 27  illustrates still another exemplary embodiment of the first tunable RF filter path  66  in the first RF filter structure  60  shown in  FIG. 21 . While the exemplary embodiment shown in  FIG. 27  is of the first tunable RF filter path  66 , any of the tunable RF filter paths shown in the first RF filter structure  60  of  FIG. 21  may be arranged in accordance with the exemplary embodiment shown in  FIG. 27 . The first tunable RF filter path  66  shown in  FIG. 27  includes the same embodiment of the resonator R( 1 , 1 ) and the same embodiment of the resonator R( 1 , 2 ) described above with respect to  FIG. 22 . In addition, the first tunable RF filter path  66  shown in  FIG. 27  includes the cross-coupling capacitive structure C(P 1 ) and the cross-coupling capacitive structure (CN 1 ) that form the V-bridge structure described above with respect to  FIG. 22 . However, the first tunable RF filter path  66  shown in  FIG. 27  further includes a resonator R( 1 , 3 ) and a resonator R( 1 , 4 ). More specifically, the resonator R( 1 , 3 ) includes an inductor  226 , a capacitive structure  228 , and a capacitive structure  230 . The resonator R( 1 , 4 ) includes an inductor  232  and a capacitive structure  234 . 
     With regard to the resonator R( 1 , 3 ), the inductor  226  is electrically connected between the capacitive structure  228  and the capacitive structure  230 . More specifically, the inductor  226  has an end  236  and an end  238 , which are disposed opposite to one another. The end  236  is electrically connected to the capacitive structure  228  and the end  238  is electrically connected to the capacitive structure  230 . Both the capacitive structure  228  and the capacitive structure  230  are grounded. Thus, the resonator R( 1 , 3 ) is a differential resonator. In this embodiment, each of the capacitive structure  228  and the capacitive structure  230  is a variable capacitive structure. 
     With regard to the resonator R( 1 , 4 ), the inductor  232  and the capacitive structure  234  are electrically connected in parallel. More specifically, the inductor  232  has an end  240  and an end  242 , which are disposed opposite to one another. The ends  240 ,  242  are each electrically connected to the capacitive structure  234 , which is grounded. Thus, the resonator R( 1 , 4 ) is a single-ended resonator. 
     In this embodiment, the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), the resonator R( 1 , 3 ), and the resonator R( 1 , 4 ) are all weakly coupled to one another. The resonator R( 1 , 3 ) and the resonator R( 1 , 4 ) are weakly coupled by providing the inductor  226  and the inductor  232  such that the inductor  226  and the inductor  232  are weakly coupled. The resonators R( 1 , 1 ), R( 1 , 2 ), R( 1 , 3 ), and R( 1 , 4 ) are each operably associated with one another such that energy transfer factors between the resonators R( 1 , 1 ), R( 1 , 2 ), R( 1 , 3 ), and R( 1 , 4 ) are less than 10%. Although the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ) are weakly coupled, the inductor  232  has a maximum lateral width and a displacement between the inductor  226  and the inductor  232  is less than or equal to half the maximum lateral width of the inductor  232 . As such, the inductor  226  and the inductor  232  are relatively close to one another. The displacement between the inductor  226  and the inductor  232  may be measured from a geometric centroid of the inductor  226  to a geometric centroid of the inductor  232 . The maximum lateral width may be a maximum dimension of the inductor  232  along a plane defined by its largest winding. The weak coupling between the inductor  226  and the inductor  232  is obtained through topological techniques. For example, the inductor  226  and the inductor  232  may be fully or partially aligned, where winding(s) of the inductor  226  and winding(s) of the inductor  232  are configured to provide weak coupling through cancellation. Alternatively or additionally, a plane defining an orientation of the windings of the inductor  226  and a plane defining an orientation of the windings of the inductor  232  may be fully or partially orthogonal to one another. 
     In some embodiments, all of the inductors  208 ,  212 ,  226 ,  232  are provided such that displacements between each of the inductors  208 ,  212 ,  226 ,  232  are less than or equal to half the maximum lateral width of the inductor  212 . Alternatively, in other embodiments, only a proper subset of the inductors  208 ,  212 ,  226 ,  232  has displacements that are less than or equal to half the maximum lateral width of the inductor  212 . For example, while the displacement between the inductor  208  and the inductor  212  may be less than or equal to half the maximum lateral width of the inductor  212  and the displacement between the inductor  226  and the inductor  232  may be less than or equal to half the maximum lateral width of the inductor  232 , the displacements from the inductor  208  and the inductor  212  to the inductor  226  and the inductor  232  may each be greater than half the maximum lateral width of the inductor  212  and half the maximum lateral width of the inductor  232 . 
     The inductors  208 ,  212 ,  226 , and  232  are magnetically coupled to the each other such that an RF signal received at the end  217  of the inductor  208  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in filtered RF signals with the same voltage polarity being transmitted out the end  220  of the inductor  212 , the end  236  of the inductor  226 , and the end  240  of the inductor  232 . Also, the inductors  208 ,  212 ,  226 , and  232  are magnetically coupled to the each other such that an RF signal received at the end  240  of the inductor  232  with a voltage polarity (i.e., either a positive voltage polarity or a negative voltage polarity) results in filtered RF signals with the same voltage polarity being transmitted out the end  217  of the inductor  208 , the end  220  of the inductor  212 , and the end  236  of the inductor  226 . This is indicated in  FIG. 27  by the dot convention where a dot is placed at the end  217  of the inductor  208 , a dot is placed at the end  220  of the inductor  212 , a dot is placed at the end  236  of the inductor  226 , and a dot is placed at the end  240  of the inductor  232 . 
     The first tunable RF filter path  66  shown in  FIG. 27  includes a cross-coupling capacitive structure C(P 3 ), a cross-coupling capacitive structure C(N 3 ), a cross-coupling capacitive structure C(P 4 ), and a cross-coupling capacitive structure C(N 4 ) electrically connected between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). With respect to the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ), the cross-coupling capacitive structure C(P 3 ), the cross-coupling capacitive structure C(N 3 ), the cross-coupling capacitive structure C(P 4 ) and the cross-coupling capacitive structure C(N 4 ) are arranged to have the X-bridge structure described above with respect to  FIG. 23 . Thus, the cross-coupling capacitive structure C(P 3 ) is electrically connected between the end  220  and the end  236  so as to provide a variable positive electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(P 3 ) is a variable cross-coupling capacitive structure configured to vary the variable positive electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). Also, the cross-coupling capacitive structure C(N 3 ) is electrically connected between the end  220  and the end  238  so as to provide a variable negative electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(N 3 ) is a variable cross-coupling capacitive structure configured to vary the variable negative electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). 
     Additionally, the cross-coupling capacitive structure C(P 4 ) is electrically connected between the end  222  and the end  238  so as to provide another variable positive electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(P 4 ) is a variable cross-coupling capacitive structure configured to vary the other variable positive electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). Finally, the cross-coupling capacitive structure C(N 4 ) is electrically connected between the end  222  and the end  236  so as to provide another variable negative electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(N 4 ) is a variable cross-coupling capacitive structure configured to vary the other variable negative electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). 
     With respect to the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ), the first tunable RF filter path  66  shown in  FIG. 27  includes a cross-coupling capacitive structure C(P 5 ) and a cross-coupling capacitive structure C(N 5 ) electrically connected between the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ). With respect to the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ), the cross-coupling capacitive structure C(P 5 ) and the cross-coupling capacitive structure C(N 5 ) are arranged to have the V-bridge structure described above with respect to  FIG. 22 . Thus, the cross-coupling capacitive structure C(P 5 ) is electrically connected between the end  236  and the end  240  so as to provide a variable positive electric coupling coefficient between the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(P 5 ) is a variable cross-coupling capacitive structure configured to vary the variable positive electric coupling coefficient provided between the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ). Also, the cross-coupling capacitive structure C(N 5 ) is electrically connected between the end  238  and the end  240  so as to provide a variable negative electric coupling coefficient between the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(N 5 ) is a variable cross-coupling capacitive structure configured to vary the variable negative electric coupling coefficient provided between the resonator R( 1 , 3 ) and the resonator R( 1 , 4 ). 
     The embodiment of first RF filter structure  60  shown in  FIG. 27  also includes a cross-coupling capacitive structure C(P 6 ), a cross-coupling capacitive structure C(N 6 ), a cross-coupling capacitive structure C(P 7 ), a cross-coupling capacitive structure C(N 7 ), and a cross-coupling capacitive structure C(P 8 ). With respect to the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ), the cross-coupling capacitive structure C(P 6 ) and the cross-coupling capacitive structure C(N 6 ) are each electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(P 6 ) is electrically connected between the end  217  and the end  236  so as to provide a variable positive electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(P 6 ) is a variable cross-coupling capacitive structure configured to vary the variable positive electric coupling coefficient provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ). Also, the cross-coupling capacitive structure C(N 6 ) is electrically connected between the end  217  and the end  238  so as to provide a variable negative electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ). The cross-coupling capacitive structure C(N 6 ) is a variable cross-coupling capacitive structure configured to vary the variable negative electric coupling coefficient provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 3 ). 
     With respect to the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ), the cross-coupling capacitive structure C(P 7 ) and the cross-coupling capacitive structure C(N 7 ) are each electrically connected between the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(P 7 ) is electrically connected between the end  220  and the end  240  so as to provide a variable positive electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(P 7 ) is a variable cross-coupling capacitive structure configured to vary the variable positive electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ). Also, the cross-coupling capacitive structure C(N 7 ) is electrically connected between the end  222  and the end  240  so as to provide a variable negative electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(N 7 ) is a variable cross-coupling capacitive structure configured to vary the variable negative electric coupling coefficient provided between the resonator R( 1 , 2 ) and the resonator R( 1 , 4 ). 
     With respect to the resonator R( 1 , 1 ) and the resonator R( 1 , 4 ), the cross-coupling capacitive structure C(P 8 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(P 8 ) is electrically connected between the end  217  and the end  240  so as to provide a variable positive electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 4 ). The cross-coupling capacitive structure C(P 8 ) is a variable cross-coupling capacitive structure configured to vary the variable positive electric coupling coefficient provided between the resonator R( 1 , 1 ) and the resonator R( 1 , 4 ). 
     Furthermore, in this embodiment, a variable capacitive structure  244  is electrically connected in series between the terminal  200  and the resonator R( 1 , 1 ). The variable capacitive structure  244  is configured to vary a variable impedance of the first tunable RF filter path  66  as measured into the terminal  200  in order to match a source or a load impedance at the terminal  200 . In addition, a variable capacitive structure  245  is electrically connected in series between the resonator R( 1 , 4 ) and the terminal  202 . The variable capacitive structure  245  is configured to vary a variable impedance of the first tunable RF filter path  66  as seen into the terminal  202  in order to match a source or a load impedance at the terminal  202 . 
       FIGS. 28A through 28D  illustrate different embodiments of the first RF filter structure  60 , wherein each of the embodiments has different combinations of input terminals and output terminals. The first RF filter structure  60  can have various topologies. For example, the embodiment of the first RF filter structure  60  shown in  FIG. 28A  has a single input terminal IN and an integer number i of output terminals OUT 1 -OUT i . As will be discussed below, the first RF filter structure  60  may define various tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  shown in  FIGS. 4, 8, 11, 12 , and  14 - 20 ) that may be used to receive different RF signals at the input terminal IN and transmit a different filtered RF signal from each of the output terminals OUT 1 -OUT i . As such, the first RF filter structure  60  shown in  FIG. 28A  may be specifically configured to provide Single Input Multiple Output (SIMO) operations. 
     With regard to the embodiment of the first RF filter structure  60  shown in  FIG. 28B , the first RF filter structure  60  has an integer number j of input terminals IN 1 -IN j  and a single output terminal OUT. As will be discussed below, the first RF filter structure  60  may define various tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  shown in  FIGS. 4, 8, 11, 12, and 14-20 ) that may be used to receive a different RF signal at each of the input terminals IN 1 -IN j  and transmit different filtered RF signals from the single output terminal OUT. As such, the first RF filter structure  60  shown in  FIG. 28B  may be specifically configured to provide Multiple Input Single Output (MISO) operations. 
     With regard to the embodiment of the first RF filter structure  60  shown in  FIG. 28C , the first RF filter structure  60  has a single input terminal IN and a single output terminal OUT. As will be discussed below, the first RF filter structure  60  may define various tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  shown in  FIGS. 4, 8, 11, 12, and 14-20 ) that may be used to receive different RF signals at the single input terminal IN and transmit different filtered RF signals from the output terminal OUT. As such, the first RF filter structure  60  shown in  FIG. 28A  may be specifically configured to provide Single Input Single Output (SISO) operations. 
     With regard to the embodiment of the first RF filter structure  60  shown in  FIG. 28D , the first RF filter structure  60  has the input terminals IN 1 -IN j  and the output terminals OUT 1 -OUT i . As will be discussed below, the first RF filter structure  60  may define various tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124  shown in  FIGS. 4, 8, 11, 12, and 14-20 ) that may be used to receive a different RF signal at each of the input terminal IN 1 -IN j  and transmit a different filtered RF signal from each of the output terminals OUT 1 -OUT i . 
       FIG. 29  illustrates another embodiment of the first RF filter structure  60 . The first RF filter structure  60  shown in  FIG. 29  includes one embodiment of the first tunable RF filter path  66  and one embodiment of the second tunable RF filter path  68 . The first tunable RF filter path  66  includes the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) are thus a first pair of weakly coupled resonators in the first tunable RF filter path  66 . The second tunable RF filter path  68  includes the resonator R( 2 , 1 ) and the resonator R( 2 , 2 ). The resonator R( 2 , 1 ) and the resonator R( 2 , 2 ) are thus a second pair of weakly coupled resonators in the second tunable RF filter path  68 . 
     As explained in further detail below, a set S of cross-coupling capacitive structures is electrically connected between the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), the resonator R( 2 , 1 ), and the resonator R( 2 , 2 ) in the first tunable RF filter path  66  and the second tunable RF filter path  68 . More specifically, the set S includes a cross-coupling capacitive structure C(PM 1 ), a cross-coupling capacitive structure C(PM 2 ), a cross-coupling capacitive structure C(PM 3 ), a cross-coupling capacitive structure C(PM 4 ), a cross-coupling capacitive structure C(NM 1 ), and a cross-coupling capacitive structure C(NM 2 ). The set S of cross-coupling capacitive structures interconnects the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), the resonator R( 2 , 1 ), and the resonator R( 2 , 2 ) so that the first RF filter structure  60  shown in  FIG. 29  is a matrix (in this embodiment, a 2×2 matrix) of the resonators R. In alternative embodiments, some of the cross-coupling capacitive structures C(PM 1 ), C(PM 2 ), C(PM 3 ), C(PM 4 ), C(NM 1 ), and C(NM 2 ) may be omitted depending on the filter transfer function to be provided. 
     Unlike in the embodiment of the first RF filter structure  60  shown in  FIG. 21 , in this embodiment, the first tunable RF filter path  66  and the second tunable RF filter path  68  are not independent of one another. The set S of cross-coupling capacitive structures thus provides for additional tunable RF filter paths to be formed from the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), the resonator R( 2 , 1 ), and the resonator R( 2 , 2 ). As discussed in further detail below, the arrangement of the first RF filter structure  60  shown in  FIG. 29  can be used to realize examples of each of the embodiments of the first RF filter structure  60  shown in  FIGS. 28A-28D . 
     The cross-coupling capacitive structure C(PM 1 ) is electrically connected within the first tunable RF filter path  66 , while the cross-coupling capacitive structure C(PM 4 ) is electrically connected within the second tunable RF filter path  68 . More specifically, the cross-coupling capacitive structure C(PM 1 ) is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) in the first tunable RF filter path  66 . The cross-coupling capacitive structure C(PM 1 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). The cross-coupling capacitive structure C(PM 4 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 2 , 1 ) and the resonator R( 2 , 2 ) in the second tunable RF filter path  68 . 
     To provide additional tunable RF filter paths, the cross-coupling capacitive structure C(PM 2 ), the cross-coupling capacitive structure C(PM 3 ), the cross-coupling capacitive structure C(NM 1 ), and the cross-coupling capacitive structure C(NM 2 ) are each electrically connected between the first tunable RF filter path  66  and the second tunable RF filter path  68 . The cross-coupling capacitive structure C(PM 2 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 2 , 2 ). The cross-coupling capacitive structure C(PM 3 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 2 , 1 ). The cross-coupling capacitive structure C(NM 1 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 1 , 1 ) and the resonator R( 2 , 2 ). The cross-coupling capacitive structure C(NM 2 ) is a variable cross-coupling capacitive structure configured to provide and vary a (e.g., positive or negative) electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 2 , 1 ). 
     The first tunable RF filter path  66  is electrically connected between the input terminal IN 1  and the output terminal OUT 1 . In addition, the second tunable RF filter path  68  is electrically connected between an input terminal IN 2  and an output terminal OUT 2 . Accordingly, the first RF filter structure  60  shown in  FIG. 29  is an embodiment of the first RF filter structure  60  shown in  FIG. 28D . However, the input terminal IN 2  and the output terminal OUT 1  are optional and may be excluded in other embodiments. For example, if the input terminal IN 2  were not provided, but the output terminal OUT 1  and the output terminal OUT 2  were provided, the first RF filter structure  60  shown in  FIG. 29  would be provided as an embodiment of the first RF filter structure  60  shown in  FIG. 28A . It might, for example, provide a diplexing or a duplexing function. Furthermore, more than two input terminals or output terminals can be provided. Some examples include embodiments of the first RF filter structure  60  used for triplexing, quadplexing, herplexing, and providing FDD and carrier aggregation. 
     The first tunable RF filter path  66  still provides a path between the input terminal IN 1  and the output terminal OUT 1 . However, assuming that the input terminal IN 2  is not provided for SIMO operation, the cross-coupling capacitive structure C(NM 1 ) is electrically connected between the first tunable RF filter path  66  and the second tunable RF filter path  68  to define a first additional tunable RF filter path between the input terminal IN 1  and the output terminal OUT 2 . The first additional tunable RF filter path is thus provided by a portion of the first tunable RF filter path  66  and a portion of the second tunable RF filter path  68 . More specifically, the first additional tunable RF filter path includes the resonator R( 1 , 1 ) and the resonator R( 2 , 2 ). The first additional tunable RF filter path also includes the cross-coupling capacitive structure C(NM 1 ) that is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ). A second additional tunable RF filter path, a third additional tunable RF filter path, a fourth additional tunable RF filter path, and a fifth additional tunable RF filter path are also defined from the input terminal IN 1  to the output terminal OUT 2 . The second additional tunable RF filter path includes the resonator R( 1 , 1 ), the cross-coupling capacitive structure C(PM 1 ), the resonator R( 1 , 2 ), the cross-coupling capacitive C(PM 2 ), and the resonator R( 2 , 2 ). Additionally, the third additional tunable RF filter path includes the resonator R( 1 , 1 ), the cross-coupling capacitive structure C(PM 3 ), the resonator R( 2 , 1 ), the cross-coupling capacitive C(PM 4 ), and the resonator R( 2 , 2 ). The fourth additional tunable RF filter path includes the resonator R( 1 , 1 ), the cross-coupling capacitive structure C(PM 1 ), the resonator R( 1 , 2 ), the cross-coupling capacitive C(NM 2 ), the resonator R( 2 , 1 ), the cross-coupling capacitive structure C(PM 4 ), and the resonator R( 2 , 2 ). Finally, the fifth additional tunable RF filter path includes the resonator R( 1 , 1 ), the cross-coupling capacitive structure C(PM 3 ), the resonator R( 2 , 1 ), the cross-coupling capacitive C(NM 2 ), the resonator R( 1 , 2 ), the cross-coupling capacitive structure C(PM 2 ), and the resonator R( 2 , 2 ). 
     If the output terminal OUT 1  were not provided, but the input terminal IN 1  and the input terminal IN 2  were provided, the first RF filter structure  60  shown in  FIG. 29  would be provided as an embodiment of the first RF filter structure  60  shown in  FIG. 28B . In this case, the second tunable RF filter path  68  still provides a path between the input terminal IN 2  and the output terminal OUT 2 . However, assuming that the output terminal OUT 1  is not provided for MISO operation, the first additional tunable RF filter path, the second additional tunable RF filter path, the third additional tunable RF filter path, the fourth additional tunable RF filter path, and the fifth additional tunable RF filter path would provide the paths from the input terminal IN 1  to the output terminal OUT 2 . 
     Finally, if the input terminal IN 2  and the output terminal OUT 2  were not provided, the first RF filter structure  60  shown in  FIG. 29  would be provided as an embodiment of the first RF filter structure  60  shown in  FIG. 28C . In this case, the second tunable RF filter path  68  still provides a path between the input terminal IN 2  and the output terminal OUT 2 . However, assuming that the output terminal IN 1  is not provided for MISO operation, the first additional tunable RF filter path, the second additional tunable RF filter path, the third additional tunable RF filter path, the fourth additional tunable RF filter path, and the fifth additional tunable RF filter path would provide the paths from the input terminal IN 1  to the output terminal OUT 2 . This may constitute a SISO filter implemented with an array to allow for a large number of signal paths and thus create one or more notches in the transfer function. 
     With regard to the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) shown in  FIG. 29 , the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) may each be single-ended resonators, differential resonators, or different combinations of single-ended resonators and differential resonators. The resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) in the first tunable RF filter path  66  may each be provided in accordance with any of the embodiments of the resonator R( 1 , 1 ) and the resonator R( 1 , 2 ) described above with respect to  FIGS. 22-27 . For example, the resonator R( 1 , 1 ) may include the inductor  208  (see  FIG. 24 ) and the capacitive structure  210  (see  FIG. 24 ). The resonator R( 1 , 2 ) may include the inductor  212  and the capacitive structure  214  (see  FIG. 24 ). The resonator R( 2 , 1 ) may include an inductor (like the inductor  208  in  FIG. 24 ) and a capacitive structure (like the capacitive structure  210  shown in  FIG. 24 ). The resonator R( 2 , 2 ) may include an inductor (like the inductor  212  in  FIG. 24 ) and a capacitive structure (like the capacitive structure  214  shown in  FIG. 24 ). 
     Additionally, one or more of the resonators R( 1 , 1 ), R( 1 , 2 ) in the first tunable RF filter path  66  and one or more of the resonators R( 2 , 1 ), R( 2 , 2 ) in the second tunable RF filter path  68  may be weakly coupled. Thus, the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) may be operably associated with one another such that an energy transfer factor between each of the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) is less than 10%. Alternatively, the energy transfer factor between only a subset of the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) is less than 10%. In addition, in at least some embodiments, not all of the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), R( 2 , 2 ) are weakly coupled to one another. 
     In this embodiment, the inductor  208  (see  FIG. 24 ) of the resonator R( 1 , 1 ), the inductor  212  (see  FIG. 24 ) of the resonator R( 1 , 2 ), the inductor of the resonator R( 2 , 1 ), and the inductor of the resonator R( 2 , 2 ) may all be weakly coupled to one another. In some embodiments, displacements between the inductor  208  (see  FIG. 24 ) of the resonator R( 1 , 1 ), the inductor  212  (see  FIG. 24 ) of the resonator R( 1 , 2 ), the inductor of the resonator R( 2 , 1 ), and the inductor of the resonator R( 2 , 2 ) may all be less than or equal to half the maximum lateral width of the inductor  212 . Alternatively, in other embodiments, only a proper subset of the inductor  208  (see  FIG. 24 ) of the resonator R( 1 , 1 ), the inductor  212  (see  FIG. 24 ) of the resonator R( 1 , 2 ), the inductor of the resonator R( 2 , 1 ), and the inductor of the resonator R( 2 , 2 ) may have displacements that are less than or equal to half the maximum lateral width of the inductor  212 . 
       FIG. 30  illustrates yet another embodiment of the first RF filter structure  60 . The first RF filter structure  60  includes the resonators R described above with respect to  FIG. 21 . The resonators R of the first RF filter structure  60  shown in  FIG. 30  are arranged as a two-dimensional matrix of the resonators R. In this embodiment, the first RF filter structure  60  includes an embodiment of the first tunable RF filter path  66 , an embodiment of the second tunable RF filter path  68 , an embodiment of the third tunable RF filter path  110 , and an embodiment of the fourth tunable RF filter path  112 . Thus, the integer M for the first RF filter structure  60  shown in  FIG. 30  is four (4) or greater. Additionally, the integer N for the first RF filter structure  60  shown in  FIG. 30  is 3 or greater. Note that in alternative embodiments, the integer M may be two (2) or greater and the integer N may be two (2) or greater. It should be noted that in alternative embodiments the number of resonators R in each row and column may be the same or different. 
     In the embodiment of the first RF filter structure  60  shown in  FIG. 30 , the first tunable RF filter path  66  includes the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), and one or more additional resonators R, such as the resonator R( 1 ,N), since the integer N is 3 or greater. All of the weakly coupled resonators R( 1 , 1 ) through R( 1 ,N) are weakly coupled to one another. Furthermore, the first tunable RF filter path  66  is electrically connected between a terminal TU 1  and a terminal TANT 1 . With regard to the second tunable RF filter path  68 , the second tunable RF filter path  68  includes the resonator R( 2 , 1 ), the resonator R( 2 , 2 ), and one or more additional resonators R, such as the resonator R( 2 ,N), since the integer N is 3 or greater. All of the weakly coupled resonators R( 2 , 1 ) through R( 2 ,N) are weakly coupled to one another. Furthermore, the second tunable RF filter path  68  is electrically connected between a terminal TU 2  and a terminal TANT 2 . 
     With regard to the third tunable RF filter path  110 , the third tunable RF filter path  110  includes a resonator R( 3 , 1 ), a resonator R( 3 , 2 ), and one or more additional resonators R, such as a resonator R( 3 ,N), since the integer N is 3 or greater. All of the weakly coupled resonators R( 3 , 1 ) through R( 3 ,N) are weakly coupled to one another. Alternatively, only a proper subset of them may be weakly coupled to one another. Furthermore, the third tunable RF filter path  110  is electrically connected between a terminal TU 3  and a terminal TANT 3 . With regard to the fourth tunable RF filter path  112 , the fourth tunable RF filter path  112  includes the resonator R(M, 1 ), the resonator R(M, 2 ), and one or more additional resonators R, such as the resonator R(M,N), since the integer N is 3 or greater. All of the weakly coupled resonators R(M, 1 ) through R(M,N) are weakly coupled to one another. Alternatively, only a proper subset of them may be weakly coupled to one another. Furthermore, the fourth tunable RF filter path  112  is electrically connected between a terminal TU 4  and a terminal TANT 4 . 
     The first tunable RF filter path  66  is configured to receive RF signals and output filtered RF signals. It should be noted that the first RF filter structure  60  may include any number of tunable RF filter paths, such as, for example, the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the fifth tunable RF filter path  122 , and the sixth tunable RF filter path  124 , described above with respect to  FIGS. 11-14 . Each of the resonators R may be a tunable resonator, which allows for a resonant frequency of each of the resonators to be varied to along a frequency range. In alternative embodiments, only a proper subset of the resonators R may be tunable. In still another embodiment, all of the resonators R are not tunable, but rather have a fixed transfer function. 
     In some embodiments, all of the resonators R in the first RF filter structure  60  shown in  FIG. 30  are weakly coupled to one another. Thus, the resonators R may all be operably associated with one another such that energy transfer factors between the resonators R are less than 10%. Alternatively, the energy transfer factor is less than 10% only among a proper subset of the resonators R. In other embodiments, only the resonators R in adjacent tunable RF filter paths  66 ,  68 ,  110 ,  112  are weakly coupled to one another. For example, all the resonators R( 1 , 1 ) through R( 1 ,N) may be weakly coupled to all the resonators R( 2 , 1 ) through R( 2 ,N). In still other embodiments, only subsets of adjacent resonators R may be weakly coupled to each other. For example, the resonators R( 1 , 1 ), R( 1 , 2 ) may be weakly coupled to the resonators R( 2 , 1 ), R( 2 , 2 ), while the resonators R( 3 , 1 ), R( 3 , 2 ) may be weakly coupled to the resonators R(M, 1 ), R(M, 2 ). These and other combinations would be apparent to one of ordinary skill in the art in light of this disclosure. 
     Sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) of cross-coupled capacitive structures are electrically connected between the resonators R. Each of the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) is arranged like the set S of cross-coupled capacitive structures described above with respect to  FIG. 29 . For example, in one particular exemplary embodiment (e.g., when M=4 and N=3), the set S( 1 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 1 , 1 ), R( 1 , 2 ) in the first tunable RF filter path  66  and the resonators R( 2 , 1 ), R( 2 , 2 ) in the second tunable RF filter path  68 . The set S( 2 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 1 , 2 ), R( 1 ,N) in the first tunable RF filter path  66  and the resonators R( 2 , 2 ), R( 2 ,N) in the second tunable RF filter path  68 . The set S( 3 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 2 , 1 ), R( 2 , 2 ) in the second tunable RF filter path  68  and the resonators R( 3 , 1 ), R( 3 , 2 ) in the third tunable RF filter path  110 . The set S( 4 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 2 , 2 ), R( 2 ,N) in the second tunable RF filter path  68  and the resonators R( 3 , 2 ), R( 3 ,N) in the third tunable RF filter path  110 . The set S( 5 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 3 , 1 ), R( 3 , 2 ) in the third tunable RF filter path  110  and the resonators R(M, 1 ), R(M, 2 ) in the fourth tunable RF filter path  112 . Finally, the set S( 6 ) of cross-coupled capacitive structures is electrically connected between the resonators R( 3 , 2 ), R( 3 ,N) in the third tunable RF filter path  110  and the resonators R(M, 2 ), R(M,N) in the fourth tunable RF filter path  112 . Note that some cross-coupled capacitive structures in the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) of cross-coupled capacitive structures for the resonators R in adjacent columns or in adjacent ones of the tunable RF filter paths  66 ,  68 ,  110 ,  112  overlap. This is because in the matrix of the resonators R, each of the resonators R is adjacent to multiple other ones of the resonators R. In another embodiment, the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) of cross-coupled capacitive structures may be connected between non-adjacent resonators R. For example, there may be cross-coupled capacitive structures between resonators R that are more than one column or row apart. 
       FIG. 31  illustrates the embodiment of the first RF filter structure  60  shown in  FIG. 30  electrically connected to the first RF antenna  16 , the second RF antenna  32 , a third RF antenna  246 , and a fourth RF antenna  247 . More specifically, the first tunable RF filter path  66  is electrically connected to the first RF antenna  16  at the terminal TANT 1 . The second tunable RF filter path  68  is electrically connected to the second RF antenna  32  at the terminal TANT 2 . The third tunable RF filter path  110  is electrically connected to the third RF antenna  246  at the terminal TANT 3 . The fourth tunable RF filter path  112  is electrically connected to the fourth RF antenna  247  at the terminal TANT 4 . With the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) of cross-coupled capacitive structures, the first RF filter structure  60  shown in  FIG. 31  forms an interconnected two-dimensional matrix of the resonators R. Thus, in addition to the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , and the fourth tunable RF filter path  112 , the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ) of cross-coupled capacitive structures provide a multitude of additional tunable RF filter paths between the terminals TU 1 , TU 2 , TU 3 , TU 4  and the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4 . It should be noted that in alternative embodiments, the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4  may not be connected to antennas. Some antennas may be omitted depending on the functionality being realized. 
     By tuning the sets S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), and S( 6 ), the first RF filter structure  60  shown in  FIG. 31  can be tuned so that any combination of the resonators R is selectable for the propagation of RF signals. More specifically, the first RF filter structure  60  shown in  FIG. 31  is tunable to route RF receive signals from any combination of the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4  to any combination of the terminals TU 1 , TU 2 , TU 3 , TU 4 . Additionally, the first RF filter structure  60  shown in  FIG. 31  is tunable to route RF transmission signals from any combination of the terminals TU 1 , TU 2 , TU 3 , TU 4  to the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4 . Accordingly, the first RF filter structure  60  can be configured to implement various MIMO, SIMO, MISO, and SISO operations. 
       FIG. 32  illustrates the first RF filter structure  60  shown in  FIGS. 30 and 31  with examples of additional tunable RF filter paths  248 ,  250  highlighted. It should be noted, however, that there are a vast number of additional combinations of the resonators R that may be selected to provide tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the additional tunable RF filter path  248 , and the additional tunable RF filter path  250 ) between the terminals TU 1 , TU 2 , TU 3 , TU 4  and the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4 . An explicit description of all of the various combinations of the resonators R that may be implemented with the first RF filter structure  60  shown in  FIGS. 30-32  is simply impractical given the high number of possible combinations. Along with the previous descriptions, the additional tunable RF filter paths  248 ,  250  are highlighted in  FIG. 32  simply to give examples of the basic concepts. However, the combinations provided for the additional tunable RF filter paths  248 ,  250  are in no way limiting, as any combination of the resonators R may be selected to route RF signals between the terminals TU 1 , TU 2 , TU 3 , TU 4  and the terminals TANT 1 , TANT 2 , TANT 3 , TANT 4 . Any number of functions, such as signal combining, splitting, multiplexing, and demultiplexing, with various filtering profiles for each, may be realized. 
     With regard to the additional tunable RF filter paths  248 ,  250  highlighted in  FIG. 32 , the additional tunable RF filter paths  248 ,  250  may be used during MIMO, SIMO, MISO, and SISO operations. More specifically, the additional tunable RF filter path  248  connects the terminal TANT 1  to the terminal TU 2 . The additional tunable RF filter path  250  connects the terminal TANT 3  to the terminal TU 2 . As such, the first RF filter structure  60  may be tuned so that the additional tunable RF filter path  248  and the additional tunable RF filter path  250  are selected in a MISO operation from the terminal TANT 1  and the terminal TANT 3  to the terminal TU 2 . The additional tunable RF filter paths  248 ,  250  may also be used in SIMO operations. For example, the first RF filter structure  60  may be tuned so that the first tunable RF filter path  66  and the additional tunable RF filter path  248  are selected in a SIMO operation from the terminal TU 2  to the terminal TANT 1 . The additional tunable RF filter paths  248 ,  250  can also be used in SISO operations from the terminal TANT 1  to the terminal TU 2  or from the terminal TANT 3  to the terminal TU 2 . Finally, the additional tunable RF filter paths  248 ,  250  may also be used in SIMO operations. For instance, the first RF filter structure  60  may be tuned so that the first tunable RF filter path  66  and the additional tunable RF filter path  250  are selected in a SIMO operation from the terminal TANT 1  to the terminal TU 1  and from the terminal TANT 3  to the terminal TU 2 . 
     In some applications involving the first RF filter structure  60  in  FIGS. 30-32 , MISO and SIMO operations can be used in conjunction with wideband antenna cables or fiber for transmitting RF signals in multiple RF communication frequency bands. Specific communication frequency bands can be processed by certain dedicated RF filtering paths in the first RF filter structure  60 . For example, different RF signals may be injected from a wideband antenna and then propagated along different dedicated tunable RF filter paths in the first RF filter structure  60  to the terminals TU 1 , TU 2 , TU 3 , TU 4 . These dedicated tunable RF filter paths can be configured to have a transfer function that is specifically designed to handle these RF signals. Furthermore, the first RF filter structure  60  shown in  FIGS. 30-32  is configured to tune a transfer function of any of the specific tunable RF filter paths (e.g., the first tunable RF filter path  66 , the second tunable RF filter path  68 , the third tunable RF filter path  110 , the fourth tunable RF filter path  112 , the additional tunable RF filter path  248 , and the additional tunable RF filter path  250 ) in the first RF filter structure  60  by tuning resonators R that are not in the specific tunable RF filter path being used to route RF signals. This can help reduce out-of-band noise and reduce insertion losses. It can also improve isolation and out-of-band attenuation. 
       FIG. 33  illustrates yet another embodiment of the first RF filter structure  60 . The first RF filter structure  60  includes the resonators R and is arranged as a two-dimensional matrix of the resonators R, where N is equal to four (4) and M is equal to three (3). In this embodiment, the first RF filter structure  60  includes an embodiment of the first tunable RF filter path  66 , an embodiment of the second tunable RF filter path  68 , and an embodiment of the third tunable RF filter path  110 . It should be noted that in alternative embodiments, the number of resonators R in each row and column may be the same or different. 
     In the embodiment of the first RF filter structure  60  shown in  FIG. 33 , the first tunable RF filter path  66  includes the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), the resonator R( 1 , 3 ), and the resonator R( 1 , 4 ). Furthermore, the first tunable RF filter path  66  is electrically connected between the terminal TU 1  and the terminal TANT 1 . With regard to the second tunable RF filter path  68 , the second tunable RF filter path  68  includes the resonator R( 2 , 1 ), the resonator R( 2 , 2 ), a resonator R( 2 , 3 ), and a resonator R( 2 , 4 ). Furthermore, the second tunable RF filter path  68  is electrically connected between the terminal TU 2  and the terminal TANT 2 . With regard to the third tunable RF filter path  110 , the third tunable RF filter path  110  includes the resonator R( 3 , 1 ), the resonator R( 3 , 2 ), a resonator R( 3 , 3 ), and a resonator R( 3 , 4 ). Furthermore, the third tunable RF filter path  110  is electrically connected between the terminal TU 3  and the terminal TANT 3 . 
     In this embodiment, the resonators R in a subset  252  of the resonators R( 1 , 1 ), R( 1 , 2 ) in the first tunable RF filter path  66  are weakly coupled to one another. A cross-coupling capacitive structure CS 1  is electrically connected between the resonators R( 1 , 1 ), R( 1 , 2 ). The cross-coupling capacitive structure CS 1  is a variable cross-coupling capacitive structure configured to vary a variable electric coupling coefficient between the resonators R( 1 , 1 ), R( 1 , 2 ). A subset  254  of the resonators R( 1 , 3 ), and R( 1 , 4 ) in the second tunable RF filter path  68  is also weakly coupled to each other. A cross-coupling capacitive structure CS 2  is electrically connected between the resonators R( 1 , 3 ), R( 1 , 4 ). The cross-coupling capacitive structure CS 2  is a variable cross-coupling capacitive structure configured to vary a variable electric coupling coefficient between the resonators R( 1 , 3 ), R( 1 , 4 ). 
     As shown in  FIG. 33 , a unidirectional coupling stage  256  is electrically connected within the first tunable RF filter path  66 . The unidirectional coupling stage  256  defines an amplifier gain and is configured to provide amplification within the first tunable RF filter path  66  in accordance with the amplifier gain. In some embodiments, the amplifier gain of the unidirectional coupling stage  256  is a variable amplifier gain. In this embodiment, the unidirectional coupling stage  256  is electrically connected between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 ). The variable amplifier gain can thus control a variable electric coupling coefficient between the resonator R( 1 , 2 ) in the subset  252  and the resonator R( 1 , 3 ) in the subset  254 . Since the unidirectional coupling stage  256  is an active semiconductor component, the unidirectional coupling stage  256  is unidirectional and thus only allows signal propagations from an input terminal IA of the unidirectional coupling stage  256  to an output terminal OA of the unidirectional coupling stage  256 . Thus, the resonator R( 1 , 2 ) in the subset  252  is unidirectionally mutual electrically coupled to the resonator R( 1 , 3 ) in the subset  254 . 
     Note that the resonators R( 1 , 3 ), R( 1 , 4 ) in the subset  254  are not electrically connected to the second tunable RF filter path  68  and the third tunable RF filter path  110 . As such, the unidirectional coupling stage  256  thus results in a portion of the first tunable RF filter path  66  with the subset  254  of the resonators R( 1 , 3 ), R( 1 , 4 ) to be unidirectional. Consequently, signal flow can be to the terminal TANT 1  but not from the terminal TANT 1 . Since the unidirectional coupling stage  256  is unidirectional, the variable amplifier gain (and thus the variable electric coupling coefficient between the resonator R( 1 , 2 ) and the resonator R( 1 , 3 )) may be controlled using feed-forward control techniques and/or feedback control techniques. 
     Next, the resonators R in a subset  258  of the resonators R( 2 , 1 ), R( 2 , 2 ), R( 3 , 1 ), and R( 3 , 2 ) in the second tunable RF filter path  68  and in the third tunable RF filter path  110  are weakly coupled to one another. An unidirectional coupling stage  260  is electrically connected between the first tunable RF filter path  66  and the second tunable RF filter path  68 . More specifically, the unidirectional coupling stage  260  is electrically connected between the resonator R( 1 , 1 ) and the resonator R( 2 , 1 ). The unidirectional coupling stage  260  defines an amplifier gain and is configured to provide amplification in accordance with the amplifier gain. In some embodiments, the amplifier gain of the unidirectional coupling stage  260  is a variable amplifier gain. The variable amplifier gain thus can control a variable electric coupling coefficient between the resonator R( 1 , 1 ) in the subset  252  and the resonator R( 2 , 1 ) in the subset  258 . A cross-coupling capacitive structure CS 3  is electrically connected between the resonator R( 1 , 2 ) and the resonator R( 2 , 2 ). The cross-coupling capacitive structure CS 3  is a variable cross-coupling capacitive structure configured to vary a variable electric coupling coefficient between the resonators R( 1 , 2 ), R( 2 , 2 ). 
     To interconnect the resonators R( 2 , 1 ), R( 2 , 2 ), R( 3 , 1 ), and R( 3 , 2 ), a set S(A) of cross-coupling capacitive structures is electrically connected between the resonators R( 2 , 1 ), R( 2 , 2 ), R( 3 , 1 ), and R( 3 , 2 ) in the subset  258 . The set S(A) of cross-coupling capacitive structures is arranged like the set S of cross-coupling capacitive structures described above with respect to  FIG. 29 . Additionally, the resonators R in a subset  262  of the resonators R( 2 , 3 ), R( 2 , 4 ), R( 3 , 3 ), and R( 3 , 4 ) in the second tunable RF filter path  68  and in the third tunable RF filter path  110  are weakly coupled to one another. A set S(B) of cross-coupling capacitive structures is electrically connected between the resonators R( 2 , 3 ), R( 2 , 4 ), R( 3 , 3 ), and R( 3 , 4 ) in the subset  262 . The set S(B) of cross-coupling capacitive structures is arranged like the set S of cross-coupling capacitive structures described above with respect to  FIG. 29 . 
     To interconnect the subset  258  and the subset  262 , the first RF filter structure  60  shown in  FIG. 33  includes a cross-coupling capacitive structure CS 4  and a unidirectional coupling stage  264 . The cross-coupling capacitive structure CS 4  is electrically connected between the resonators R( 2 , 2 ), R( 2 , 3 ). The cross-coupling capacitive structure CS 4  is a variable cross-coupling capacitive structure configured to vary a variable electric coupling coefficient between the resonators R( 2 , 2 ), R( 2 , 3 ). The unidirectional coupling stage  264  is electrically connected within the third tunable RF filter path  110 . In this embodiment, the unidirectional coupling stage  264  is electrically connected between the resonator R( 3 , 3 ) and the resonator R( 3 , 2 ). The unidirectional coupling stage  264  defines an amplifier gain and is configured to provide amplification within the third tunable RF filter path  110  in accordance with the amplifier gain. In some embodiments, the amplifier gain of the unidirectional coupling stage  264  is a variable amplifier gain. The variable amplifier gain can thus control a variable electric coupling coefficient between the resonator R( 3 , 3 ) in the subset  262  and the resonator R( 3 , 2 ) in the subset  258 . Since the unidirectional coupling stage  264  is an active semiconductor component, the unidirectional coupling stage  264  is unidirectional and thus only allows signal propagations from an input terminal IB of the unidirectional coupling stage  264  to an output terminal OB of the unidirectional coupling stage  264 . Thus, the resonator R( 3 , 3 ) in the subset  262  is unidirectionally mutual electrically coupled to the resonator R( 3 , 2 ) in the subset  258 . Consequently, the third tunable RF filter path  110  shown in  FIG. 33  is unidirectional if the signal flow is between the terminal TANT 3  and the terminal TU 3  though the third tunable RF filter path  110 . As such signal flow between the terminal TANT 3  and the terminal TU 3  is provided only through the third tunable RF filter path  110 , signal flow can only be from the terminal TANT 3  to the terminal TU 3 , and not vice versa. In other cases, an additional tunable RF signal path (e.g., the additional RF terminal tunable RF signal path that includes the resonators R( 3 , 1 ), R( 2 , 2 ), R( 2 , 3 ) and R( 3 , 4 )) can be tuned to provide bidirectional signal flow between the terminal TU 3  and the terminal TANT 3  through the cross-coupling capacitive structure CS 4 . The unidirectional coupling stages  256 ,  260 ,  264  may be active devices, such as amplifiers, diodes, transistors, networks of transistors, buffer stages, attenuation stages, and the like. The unidirectional coupling stages  256 ,  260 ,  264  can have gains higher than one (1), lower than one (1), or equal to one (1). Additionally, the unidirectional coupling stages  256 ,  260 ,  264  may be passive devices. The unidirectional coupling stages  256 ,  260 ,  264  may not be entirely or ideally unilateral, but may have some finite reverse coupling. In this case, the unidirectional coupling stages  256 ,  260 ,  264  may be predominately unilateral. One example in which the unidirectional coupling stages  256 ,  260 ,  264  may be used for multi-resonator applications and may improve isolation between certain parts, such as transmission ports and receive ports of a duplexer. 
       FIG. 34  illustrates yet another embodiment of the first RF filter structure  60 . The first RF filter structure  60  shown in  FIG. 34  is integrated into an IC package  266 . The first RF filter structure  60  shown in  FIG. 34  includes the resonators R and is arranged as a two-dimensional matrix of the resonators R, where N is equal to three (3) and M is equal to two (2). It should be noted that in alternative embodiments the number of resonators R in each row and column may be the same or different. 
     In this embodiment, the first RF filter structure  60  includes an embodiment of the first tunable RF filter path  66  and an embodiment of the second tunable RF filter path  68 . The first tunable RF filter path  66  includes the resonator R( 1 , 1 ), the resonator R( 1 , 2 ), and the resonator R( 1 , 3 ). The second tunable RF filter path  68  includes the resonator R( 2 , 1 ), the resonator R( 2 , 2 ), and the resonator R( 2 , 3 ). A set S(X) of cross-coupling capacitive structures is electrically connected between the resonators R( 1 , 1 ), R( 1 , 2 ), R( 2 , 1 ), and R( 2 , 2 ). The set S(X) of cross-coupling capacitive structures is arranged like the set S of cross-coupling capacitive structures described above with respect to  FIG. 29 . A set S(Y) of cross-coupling capacitive structures is electrically connected between the resonators R( 1 , 2 ), R( 1 , 3 ), R( 2 , 2 ), and R( 2 , 3 ). The set S(Y) of cross-coupling capacitive structures is also arranged like the set S of cross-coupling capacitive structures described above with respect to  FIG. 29 . 
     As shown in  FIG. 34 , the IC package  266  houses a package substrate  268 , a semiconductor die  270 , and a semiconductor die  272 . The semiconductor die  270  and the semiconductor die  272  are mounted on the package substrate  268 . In this embodiment, the resonators R of the first RF filter structure  60  are formed by the package substrate  268 . The set S(X) of cross-coupling capacitive structures is formed by the semiconductor die  270 . On the other hand, the set S(Y) of cross-coupling capacitive structures is formed by the semiconductor die  272 . Thus, the set S(X) of cross-coupling capacitive structures and the set S(Y) of cross-coupling capacitive structures are formed on multiple and separate semiconductor dies  270 ,  272 . Using the multiple and separate semiconductor dies  270 ,  272  may be helpful in order to increase isolation. The multiple and separate semiconductor dies  270 ,  272  may have less area than the semiconductor die  268  shown in  FIG. 34 . As such, the embodiment shown in  FIG. 35  may consume less die area. 
       FIG. 35  illustrates another embodiment of an IC package  266 ′ that houses the same embodiment of the first RF filter structure  60  described above with regard to  FIG. 34 . The IC package  266 ′ is the same as the IC package  266  shown in  FIG. 34 , except that the IC package  266 ′ only has a single semiconductor die  274 . In this embodiment, both the set S(X) of cross-coupling capacitive structures and the set S(Y) of cross-coupling capacitive structures are formed by the semiconductor die  272 . Thus, the IC package  266 ′ allows for a more compact arrangement than the IC package  266 . 
       FIG. 36  illustrates yet another embodiment of the first RF filter structure  60 . In this embodiment, the first RF filter structure  60  is arranged as a three-dimensional matrix of resonators R 1 , R 2 , R 3 . More specifically, a two-dimensional matrix of the resonators R 1  is provided on a plane k, a two-dimensional array of the resonators R 2  is provided on a plane m, and a two-dimensional array of the resonators R 3  is provided on a plane n. Cross-coupling capacitive structures CC are electrically connected between the resonators R 1 , R 2 , R 3  that are adjacent to one another in the same plane k,m,n and in the different planes k,m,n. The three-dimensional matrix of resonators R 1 , R 2 , R 3  thus allows for more resonators to be cross-coupled to one another. This allows for the first RF filter structure  60  to provide greater numbers of tunable RF filter paths and allows for the first RF filter structure  60  to be tuned more accurately. 
     In general, having more tunable RF filter paths allows for the synthesis of a more complex transfer function with multiple notches for better blocker rejection. The number of resonators R 1 , R 2 , R 3  in each of the planes k, n, m may be different or the same. The three-dimensional matrix of resonators can be used in MIMO, SIMO, MISO, and SISO applications. 
       FIG. 37  illustrates an exemplary RF device  300 . For example, the RF device  300  may be or may include the embodiments of the first RF filter structure  60  shown in  FIGS. 4, 7, 8, 11-21, 28A-28D, and 29-36 . In this embodiment, the RF device  300  is integrated into one embodiment of an integrated circuit (IC) package  302  (e.g., the IC package  266  shown in  FIG. 35  or the IC package  266 ′ shown in  FIG. 36 ) that includes a semiconductor die  304  (e.g., the semiconductor die  270  shown in  FIG. 34 , the semiconductor die  272  shown in  FIG. 34 , and the semiconductor die  274  shown in  FIG. 35 ) and a package board  306  (e.g., the package substrate  268  shown in  FIGS. 34 and 35 ). The RF device  300  includes a resonator (e.g., the resonators R, R 1 , R 2 , and R 3  described above in  FIGS. 21-27  and  FIGS. 29-36 ) that includes an inductor  308  (e.g., the inductor  208  illustrated in  FIGS. 21-27 , the inductor  212  illustrated in  FIGS. 21-27 , the inductor  226  illustrated in  FIG. 27 , and the inductor  232  illustrated in  FIG. 27 ), a capacitive structure  310 A (e.g., any or part of the capacitive structures  210 ,  214 ,  216 ,  224 ,  228 ,  230 ,  234 , any or part of the cross-coupled capacitive structures C, C(P 1 ), C(P 2 ), C(P 3 ), C(P 4 ), C(P 5 ), C(P 6 ), C(P 7 ), C(P 8 ), C(N 1 ), C(N 2 ), C(N 3 ), C(N 4 ), C(N 5 ), C(N 6 ), C(N 7 ), C(PH 1 ), C(PH 2 ), C(PH 3 ), C(PH 4 ), C(NH 1 ), C(NH 2 ), C(NH 3 ), C(NH 4 ), C(I 1 ), C(I 2 ), CS 1 , CS 2 , CS 3 , CS 4 , or any or part of the cross-coupled capacitive structures in the sets S, S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), S( 6 ), S(A), S(B), S(X), and S(Y) shown in  FIGS. 21-27 and 29-36 ), and a capacitive structure  310 B (e.g., any or part of the capacitive structures  210 ,  214 ,  216 ,  224 ,  228 ,  230 ,  234 , any or part of the cross-coupled capacitive structures C, C(P 1 ), C(P 2 ), C(P 3 ), C(P 4 ), C(P 5 ), C(P 6 ), C(P 7 ), C(P 8 ), C(N 1 ), C(N 2 ), C(N 3 ), C(N 4 ), C(N 5 ), C(N 6 ), C(N 7 ), C(PH 1 ), C(PH 2 ), C(PH 3 ), C(PH 4 ), C(NH 1 ), C(NH 2 ), C(NH 3 ), C(NH 4 ), C(I 1 ), C(I 2 ), CS 1 , CS 2 , CS 3 , CS 4 , or any or part of the cross-coupled capacitive structures in the sets S, S( 1 ), S( 2 ), S( 3 ), S( 4 ), S( 5 ), S( 6 ), S(A), S(B), S(X), and S(Y) shown in  FIGS. 21-27 and 29-36 ). (It should be noted that the capacitive structures  310 A,  310 B are referred to generically as element  310 ). 
     The inductor  308  may be provided in the resonator (e.g., the resonators R, R 1 , R 2 , and R 3  described with regard to  FIGS. 21-27 and 29-36 ) and the capacitive structures  310 A,  310 B may be part of or may be electrically connected to the resonator (e.g., the resonators R, R 1 , R 2 , and R 3  shown in  FIGS. 21-27 and 29-36 ) and should have a high Q factor in order to optimize performance and reduce noise distortion. In many applications, however, the inductors (e.g., the inductor  308 ) and the capacitive structures (e.g. the capacitive structures  310 A,  310 B) used in or electrically connected to the resonators are provided on a combination of different electronic support architectures, such as semiconductor dies, package boards for IC packages, printed circuit boards (PCBs), or any combination of the like. For example, in the embodiment shown in  FIG. 37 , the inductor  308  may be formed by the package board  306  while the capacitive structures  310 A,  310 B are formed by the semiconductor die  304 . 
     As explained in further detail below, the RF device  300  shown in  FIG. 37  allows for interconnections between the inductor  308  in the package board  306 , the capacitive structures  310 A,  310 B, and other components in the semiconductor die  304  without significantly degrading the high Q factor of the resonator. More specifically, the interconnections provided by the RF device  300  prevent or reduce current crowding, which significantly reduces a Q factor of components in RF devices (not shown) of previously known structures. While the interconnections provided here are provided with the inductor  308  formed by one electronic support architecture (e.g., the package board  306 ) and the capacitive structures  310  are formed by another electronic support architecture (e.g., the semiconductor die  304 ), the techniques described herein to provide interconnections are equally applicable when the interconnections need to be provided and the inductor  308  and the capacitive structures  310  are formed on a single electronic support architecture (e.g., a single multi-layered substrate). These and other applications would be apparent to one of ordinary skill in the art in light of this disclosure. 
     The inductor  308  has an inductor terminal  312 A and an inductor terminal  312 B. At the inductor terminals  312 A,  312 B, the inductor  308  is configured to receive and/or transmit a current  313 . The current  313  is shown propagating from the inductor terminal  312 A, through the capacitive structures  310 A,  310 B in the semiconductor die  304  and to the inductor terminal  312 B. The semiconductor die  304  may include one or more active semiconductor devices (referred to generically as elements  314  and specifically as elements  314 A- 314 N). The semiconductor die  304  includes N number of the active semiconductor devices  314 , where N is an integer greater than or equal to one. In this embodiment, the active semiconductor devices  314  are a stack of field-effect transistors (FETs). However, each of the active semiconductor devices  314  may be any type of active semiconductor device, such as other types of transistors, diodes, metal-on-semiconductor (MOS) capacitive structures, varactors, on-chip resistive structures, metal-insulator-metal (MIM) capacitive structures, metal-on-metal capacitive structures, and/or the like. As shown in  FIG. 37 , the active semiconductor devices  314  include a device contact  316 A and a device contact  316 B. In this embodiment, the device contact  316 A is a drain contact of the active semiconductor device  314 A at a first end of the stack of FETs and the device contact  316 B is a source contact of the active semiconductor device  314 B at a second end of the stack of FETs opposite the first end. The active semiconductor devices  314  thus electrically connect the device contact  316 A to the device contact  316 B. In this manner, the active semiconductor devices  314  form a switch path between the device contact  316 A and the device contact  316 B. 
     As shown in  FIG. 37 , the device contact  316 A is positioned so as to be vertically aligned directly below the inductor terminal  312 A. Additionally, the device contact  316 B is positioned so as to be vertically aligned directly below the inductor terminal  312 B. To interconnect the inductor terminal  312 A to the device contact  316 A, the semiconductor die  304  includes an interconnection path  318 A that electrically connects the inductor terminal  312 A to the device contact  316 A. The capacitive structure  310 A is formed within the interconnection path  318 A. Similarly, the semiconductor die  304  includes an interconnection path  318 B that electrically connects the inductor terminal  312 B to the device contact  316 B to interconnect the inductor terminal  312 B to the device contact  316 B. Since the active semiconductor devices  314  are stacked, the active semiconductor devices  314  provide the switch path that electrically connects the interconnection path  318 A to the interconnection path  318 B so that the interconnection path  318 A, the switch path, and the interconnection path  318 B provide a current path between the inductor terminal  312 A and the inductor terminal  312 B. It should be noted that in some embodiments the inductor terminal  312 A is at one end of the inductor  308  while the inductor terminal  312 B is at another end of the inductor  308 . However, the inductor terminal  312 A may also be a conductive pad that is electrically connected through a trace or the like to one end of the inductor  308 . Similarly, the inductor terminal  312 B may be a conductive pad that is electrically connected through a trace or the like to another end of the inductor  308 . 
     The active semiconductor devices  314  are configured to be turned on and turned off. When the active semiconductor devices  314  are turned on, the active semiconductor devices  314  are configured to close the current path for the current  313 . The current  313  thus can propagate through the current path from the inductor terminal  312 A to the inductor terminal  312 B. As such, a capacitance of the capacitive structure  310 A and a capacitance of the capacitive structure  310 B are presented between the inductor terminals  312 A,  312 B. When the active semiconductor devices  314  are turned off, the active semiconductor devices  314  are configured to open the current path for the current  313 . The current  313  cannot propagate from or to the inductor terminals  312 A,  312 B though the current path. Thus, the capacitance of the capacitive structure  310 A and the capacitance of the capacitive structure  310 B are not presented between the inductor terminals  312 A,  312 B, since the current path appears as an open circuit. In some implementations, there may be a finite parasitic capacitance between the inductor terminals  312 A,  312 B, even when the active semiconductor devices  314  are turned off. 
     To help maintain a high Q factor, the interconnection path  318 A is vertically aligned so as to extend directly between the inductor terminal  312 A and the device contact  316 A. Similarly, the interconnection path  318 B is vertically aligned so as to extend directly between the inductor terminal  312 B and the device contact  316 B. By vertically aligning the device contact  316 A directly below the inductor terminal  312 A and the device contact  316 B directly below the inductor terminal  312 B, the current  313  does not have to be routed back and forth vertically and horizontally in order to propagate from the inductor terminal  312 A to the device contact  316 A and from the device contact  316 B to the inductor terminal  312 B. This prevents additional losses and current crowding, since the current  313  does not have to go back and forth through metallic layers of different widths and thicknesses. 
     With regard to the interconnection path  318 A, the inductor terminal  312 A has an inductor terminal connection surface  320 A that is attached to the interconnection path  318 A, while the device contact  316 A has a device contact connection surface  322 A that is attached to the interconnection path  318 A. The interconnection path  318 A is formed to extend vertically such that the interconnection path  318 A is substantially orthogonal to both the inductor terminal connection surface  320 A and the device contact connection surface  322 A. The interconnection path  318 A extends directly between the inductor terminal  312 A and the device contact  316 A such that the current  313  has a direction D 1  of current flow through the interconnection path  318 A that is substantially orthogonal to a direction D 2  of current flow of the current  313  through the inductor terminal  312 A. As shown in  FIG. 37 , the entire interconnection path  318 A is aligned directly between the inductor terminal connection surface  320 A and the device contact connection surface  322 A. Furthermore, the interconnection path  318 A extends directly between the inductor terminal  312 A and the device contact  316 A such that the direction D 1  of current flow through the interconnection path  318 A is also substantially orthogonal to a direction D 3  of current flow of the current  313  through the active semiconductor devices  314 . 
     With regard to the interconnection path  318 B, the inductor terminal  312 B has an inductor terminal connection surface  320 B that is attached to the interconnection path  318 B while the device contact  316 B has a device contact connection surface  322 B that is attached to the interconnection path  318 B. The interconnection path  318 B is formed to extend vertically such that the interconnection path  318 B is substantially orthogonal to both the inductor terminal connection surface  320 B and the device contact connection surface  322 B. The interconnection path  318 B extends directly between the inductor terminal  312 B and device contact  316 B such that the current  313  has a direction D 4  of current flow through the interconnection path  318 B that is substantially orthogonal to a direction D 5  of current flow of the current  313  through the inductor terminal  312 B. As shown in  FIG. 37 , the entire interconnection path  318 B is aligned directly between the inductor terminal connection surface  320 B and the device contact connection surface  322 B. To minimize a length of the current path and thus also insertion losses, the interconnection path  318 B extends directly between the inductor terminal  312 B and the device contact  316 B such that the direction D 4  of current flow through the interconnection path  318 B is also substantially orthogonal to the direction D 3  of current flow of the current  313  through the active semiconductor devices  314 . As shown in  FIG. 37 , the direction D 4  through the interconnection path  318 B is antipodal to the direction D 2  through the interconnection path  318 A. Furthermore, a first horizontal displacement between the inductor terminal  312 A and the inductor terminal  312 B is substantially equal to a second horizontal displacement between the device contact  316 A and the device contact  316 B. 
     In the embodiment shown in  FIG. 37 , the inductor  308  is formed by the package board  306 . The package board  306  may have a package body  324  and metallic layers formed on or in the package body  324 . The package body  324  may be formed by a plurality of board layers made from a non-conductive material. The non-conductive material may be a dielectric, a laminate, fibers, glass, ceramic, and/or the like. The dielectric may be a Silicon Oxide (SiOx), Silicon Nitride (SiNx), and/or the like. In this embodiment, the package board  306  is a laminate. The laminate may be FR-1, FR-2, FR-3, FR-4, FR-5, FR-6, CEM-1, CEM-2, CEM-3, CEM-4, CEM-5, CX-5, CX-10, CX-20, CX-30, CX-40, CX-50, CX-60, CX-70, CX-80, CX-90, CX-100, and/or the like. The metallic layers of the package board  306  may be used to form termini and passive impedance components (e.g., the inductor  308 ). 
     In this particular embodiment, the package board  306  is a laminate that defines a metallic layer  326  that is formed on a laminate surface  328 . The metallic layer  326  is used to form the inductor terminals  312 A,  312 B of the inductor  308 . The inductor  308  may be any type of suitable inductor. For example, the inductor  308  may be a folded inductor. Additionally, the inductor  308  may be a two-dimensional (2D) inductor or three-dimensional (3D) inductor. If the inductor  308  is the 2D inductor, the inductor  308  may be formed entirely from the metallic layer  326 . On the other hand, if the inductor  308  is a 3D inductor, then multiple metallic layers provided by the package board  306  may be used to form the inductor  308 . The current flow in the inductor  308  may be directed both horizontally and vertically, as in the case with 3D inductors, or just horizontally, as in the case of standard planar horizontal inductors. Additionally, the inductor  308  may be a vertical inductor. Thus, the current flow in the inductor  308  may also only be vertical. 
     As shown in  FIG. 37 , the semiconductor die  304  is attached to the package board  306 . The semiconductor die  304  includes a semiconductor substrate  329  used to form active semiconductor components of the IC. More specifically, the active semiconductor devices  314  are formed by the semiconductor substrate  329 . The semiconductor substrate  329  may be formed from doped and non-doped layers of a suitable semiconductor material. For example, the semiconductor material may be Silicon (Si), Silicon Germanium (SiGe), Gallium Arsenide (GaAs), Indium Phosphorus (InP), and/or the like. Typical dopants that may be utilized to dope the semiconductor layers are Gallium (Ga), Arsenic (As), Silicon (Si), Tellurium (Te), Zinc (Zn), Sulfur (S), Boron (B), Phosphorus (P), Aluminum Gallium Arsenide (AlGaAs), Indium Gallium Arsenide (InGaAs), and/or the like. Furthermore, metallic layers may be formed on a top, within, and/or on a bottom of the semiconductor substrate  329  to provide termini of the active semiconductor components, to form passive impedance elements, and/or the like. For example, a metallic layer is used to form the device contacts  316 A,  316 B of the active semiconductor devices  314 . Insulating layers, such as oxide layers, and metallic layers may also be provided in or on the semiconductor substrate  329 . 
     The semiconductor die  304  also includes a Back-End-of-Line (BEOL)  330  that is formed over the semiconductor substrate  329 . The BEOL  330  may be formed from a non-conductive substrate and a plurality of metallic layers provided on or in the insulating substrate. As shown in  FIG. 37 , the interconnection path  318 A that electrically connects the inductor terminal  312 A to the device contact  316 A is formed by the BEOL  330 . Since the capacitive structure  310 A is formed within the interconnection path  318 A, the BEOL  330  also forms the capacitive structure  310 A. Additionally, the interconnection path  318 B that electrically connects the inductor terminal  312 B to the device contact  316 B is also formed by the BEOL  330 . Since the capacitive structure  310 B is formed within the interconnection path  318 B, the BEOL  330  also forms the capacitive structure  310 B. 
     The BEOL  330  includes a metallic layer M 1 , metallic layer M 2 , a metallic layer M 3 , and a metallic layer M 4 . Conductive vias are also provided in the BEOL  330  to couple the components on the semiconductor substrate  329  to one another. In this embodiment, the interconnection path  318 A is formed by the metallic layers M 1 , M 2 , M 3 , and M 4 , and by conductive vias VA that interconnect the metallic layers M 1 , M 2 , M 3 , and M 4  within the interconnection path  318 A. Note that the metallic layer M 1  is used to form a conductive pad in the interconnection path  318 A that allows for external connections to the semiconductor die  304 . This conductive pad is vertically aligned directly over the device contact  316 A. In this embodiment, a connection pillar CPA electrically connects the inductor terminal  312 A to the conductive pad formed by the metallic layer M 1  in the interconnection path  318 A. In fact, all of the metallic components in the interconnection path  318 A are vertically aligned directly over the device contact  316 A. 
     The interconnection path  318 A has a path width PWA and the inductor terminal  312 A has a line width LWA. To reduce and/or prevent current crowding within the interconnection path  318 A, the path width PWA and the line width LWA are substantially equal. With regard to the capacitive structure  310 A shown in  FIG. 37 , a top plate is formed from the metallic layer M 2  and a bottom plate is formed from the metallic layer M 3 . The interconnection path  318 A also includes a dielectric material DMA between the top plate and the bottom plate. The capacitive structure  310 A is vertically aligned directly over the device contact  316 A and directly underneath the inductor terminal  312 A. Furthermore, the path width PWA of the interconnection path  318 A is maintained substantially constant throughout the entire interconnection path  318 A. As such, a top plate width of the top plate, a bottom plate width of the bottom plate, and the line width LWA are substantially equal. In this manner, the capacitive structure  310 A is configured within the interconnection path  318 A to prevent or reduce current crowding. 
     Also, in this embodiment, the interconnection path  318 B is formed by the metallic layers M 1 , M 2 , M 3 , and M 4 , and by conductive vias VB that interconnect the metallic layers M 1 , M 2 , M 3 , and M 4  within the interconnection path  318 B. Note that the metallic layer M 1  is used to form a conductive pad that is in the interconnection path  318 B and allows for external connections to the semiconductor die  304 . This conductive pad is vertically aligned directly over the device contact  316 B. In this embodiment, a connection pillar CPB electrically connects the inductor terminal  312 B to the conductive pad formed by the metallic layer M 1  in the interconnection path  318 B. In fact, all of the metallic components in the interconnection path  318 B are vertically aligned directly over the device contact  316 B. In alternative embodiments, the inductor  308  may formed by the BEOL  330 . For example, the BEOL  330  may include several thick metallic layers. One or more of these thick metallic layers may be used to form the inductor  308 . In this manner, the inductor  308  may be provided by the semiconductor die  304 , while still providing a high Q factor. 
     The interconnection path  318 B has a path width PWB and the inductor terminal  312 B has a line width LWB. To reduce and/or prevent current crowding within the interconnection path  318 B, the path width PWB and the line width LWB are substantially equal. With regard to the capacitive structure  310 B shown in  FIG. 37 , a top plate is formed from the metallic layer M 2  and a bottom plate is formed from the metallic layer M 3 . It should be noted that other metallic layers may be used to form the top plate and the bottom plate. The interconnection path  318 B also includes a dielectric material DMB between the top plate and the bottom plate. The capacitive structure  310 B is vertically aligned directly over the device contact  316 B and directly underneath the inductor terminal  312 B. Furthermore, the path width PWB of the interconnection path  318 B is maintained substantially constant throughout the entire interconnection path  318 B. As such, a top plate width of the top plate, a bottom plate width of the bottom plate, and the line width LWB are substantially equal. Accordingly, the capacitive structure  310 B is configured within the interconnection path  318 B to prevent or reduce current crowding and minimize the length of the current path. If an area of the capacitive structure  310 A and an area of the capacitive structure  310 B are each smaller than an area of the inductor terminal  312 A and the inductor terminal  312 B, respectively, the capacitive structures  310 A,  310 B may be split into smaller capacitive structures and distributed uniformly within the interconnection paths  318 A,  318 B. The same is true if the area of the capacitive structures  310 A,  310 B is smaller than an area of the interconnection paths  318 A,  318 B. 
       FIG. 38  illustrates another embodiment of the RF device  300  having the interconnection path  318 A that electrically connects the inductor terminal  312 A to device contact  316 A, as described above in  FIG. 37 . However, in this embodiment, the number N is equal to one, and thus the RF device  300  shown in  FIG. 38  just has the active semiconductor device  314 A. Also, the interconnection path  318 B (shown in  FIG. 37 ) is not provided. Thus, the RF device  300  shown in  FIG. 38  only includes the active semiconductor device  314 A with the device contact  316 A and the interconnection path  318 A. Furthermore, a device contact  332 A of the active semiconductor device  314 A is grounded. For example, the active semiconductor device  314 A may be grounded to a real ground or a virtual ground. Thus, the active semiconductor device  314 A and the interconnection path  318 A provide a current path between the inductor terminal  312 A and ground. In this embodiment, the active semiconductor device  314 A is a FET and the device contact  316 A is a drain contact that is electrically connected to a drain DA of the active semiconductor device  314 A. The device contact  332 A is a source contact electrically connected to a source SA of the active semiconductor device  314 A. (As shown in  FIG. 37  multiple active semiconductor devices  314  may also form a stack of active semiconductor devices  314 .) 
     The interconnection path  318 A and the active semiconductor device  314 A thus provide a current path for the current  313  from the inductor terminal  312 A to ground. In this embodiment, the active semiconductor device  314 A also includes a control contact  334 A configured to receive a control voltage V CA . Since the active semiconductor device  314 A is provided as the FET, the control contact  334 A shown in  FIG. 38  is a gate contact that electrically connects to a gate GA of the active semiconductor device  314 A. The active semiconductor device  314 A is configured to close the current path from the inductor terminal  312 A to ground in response to the control voltage V CA  being provided in an activation voltage state. In this case, the current  313  propagates from the inductor terminal  312 A provided by the package board  306  through the interconnection path  318 A and the active semiconductor device  314 A to ground. As such, the capacitance of the capacitive structure  310 A is presented at the inductor terminal  312 A when the active semiconductor device  314 A is turned on. The active semiconductor device  314 A is configured to open the current path from the inductor terminal  312 A to ground in response to the control voltage V CA  being provided in a deactivation voltage state. Thus, the current  313  does not propagate to ground and the capacitance of the capacitive structure  310 A is not presented at the inductor terminal  312 A when the active semiconductor device  314 A is turned off. In some embodiments, parasitic capacitances may provide a parasitic current path. 
     Since the interconnection path  318 A is vertically aligned so as to extend directly between the inductor terminal  312 A and the device contact  316 A, a length of the interconnection path  318 A is minimized. Thus, parasitic impedances presented by the interconnection path  318 A are minimized. For instance, a parasitic inductance and a parasitic resistance of the interconnection path  318 A are minimized. Note that in this embodiment, a solder ball SBA electrically connects the interconnection path  318 A to the inductor terminal  312 A, rather than the conductive pillar CPA shown in  FIG. 37 . The current path may have a close contour with a certain path length. 
       FIG. 39  illustrates another embodiment of the IC package  302  with an embodiment of an RF device  300 ( 1 ) and an embodiment of an RF device  300 ( 2 ) integrated into the IC package  302 . Both the RF device  300 ( 1 ) and the RF device  300 ( 2 ) are similar to the RF device  300  shown in  FIG. 37 , except that the integer number N for both the RF device  300 ( 1 ) and the RF device  300 ( 2 ) is equal to one (1). As shown in  FIG. 39 , the RF device  300 ( 1 ) includes a capacitive structure  310 A( 1 ), a capacitive structure  310 B( 1 ), an inductor terminal  312 A( 1 ), an inductor terminal  312 B( 1 ), an active semiconductor device  314 A( 1 ) that includes a device contact  316 A( 1 ) and a device contact  332 A( 1 ), an interconnection path  318 A( 1 ), and an interconnection path  318 B( 1 ). The capacitive structure  310 A( 1 ), the capacitive structure  310 B( 1 ), the inductor terminal  312 A( 1 ), the inductor terminal  312 B( 1 ), the active semiconductor device  314 A( 1 ), the interconnection path  318 A( 1 ), and the interconnection path  318 B( 1 ) are the same as the capacitive structure  310 A, the capacitive structure  310 B, the inductor terminal  312 A, the inductor terminal  312 B, the active semiconductor device  314 A, the interconnection path  318 A, and the interconnection path  318 B shown in  FIG. 37 . However, since the integer number N for the RF device  300 ( 1 ) is equal to one (1), the interconnection path  318 B( 1 ) electrically connects the inductor terminal  312 B( 1 ) to the device contact  332 A( 1 ) of the active semiconductor device  314 A( 1 ). 
     With regard to the RF device  300 ( 2 ), the RF device  300 ( 2 ) includes a capacitive structure  310 A( 2 ), a capacitive structure  310 B( 2 ), an inductor terminal  312 A( 2 ), an inductor terminal  312 B( 2 ), an active semiconductor device  314 A( 2 ) that includes a device contact  316 A( 2 ) and a device contact  332 A( 2 ), an interconnection path  318 A( 2 ), and an interconnection path  318 B( 2 ). The capacitive structure  310 A( 2 ), the capacitive structure  310 B( 2 ), the inductor terminal  312 A( 2 ), the inductor terminal  312 B( 2 ), the active semiconductor device  314 A( 2 ), the interconnection path  318 A( 2 ), and the interconnection path  318 B( 2 ) are the same as the capacitive structure  310 A, the capacitive structure  310 B, the inductor terminal  312 A, the inductor terminal  312 B, the active semiconductor device  314 A, the interconnection path  318 A, and the interconnection path  318 B shown in  FIG. 37 . However, since the integer number N for the RF device  300 ( 2 ) is equal to one (1), the interconnection path  318 B( 2 ) electrically connects the inductor terminal  312 B( 2 ) to the device contact  332 A( 2 ) of the active semiconductor device  314 A( 2 ). 
     As shown in  FIG. 39 , the inductor terminal  312 B( 1 ) of the RF device  300 ( 1 ) is electrically connected to the inductor terminal  312 A( 1 ) through a trace  336  provided by the package board  306 . As shown in  FIG. 39 , a configuration of the RF devices  300 ( 1 ) and  300 ( 2 ) allows for connections between the RF device  300 ( 1 ) and the RF device  300 ( 2 ) using thicker metallic layers, such as the metallic layer that forms the trace  336  and the inductor terminals  312 A( 1 ),  312 A( 2 ),  312 B( 1 ), and  312 B( 2 ). The metallic layers M 2 , M 3 , and M 4  within the BEOL  330  may be thin in comparison to the metallic layer M 1  at a surface of the BEOL  330  and the metallic layers in the package board  306 . Thus, the configuration of the RF devices  300 ( 1 ),  300 ( 2 ) routes the current vertically through the interconnection paths  318 A( 1 ),  318 B( 1 ),  318 A( 2 ),  318 B( 2 ) to and from the active semiconductor devices  314 A( 1 ),  314 A( 2 ). In order to route the current  313  between the RF devices  300 ( 1 ),  300 ( 2 ), the trace  336  is horizontally oriented. However, since the trace  336  is relatively thick, current crowding is reduced or prevented, and thus a Q factor of the RF devices  300 ( 1 ),  300 ( 2 ) is not degraded. In alternative embodiments, the trace  336  may be realized by a thick metal layer in the BEOL  330 . 
       FIG. 40  illustrates another embodiment of the IC package  302  with another embodiment of an RF device  300 ( 1 ) and another embodiment of the RF device  300 ( 2 ) integrated into the IC package  302 . In the embodiment of the RF device  300 ( 1 ) shown in  FIG. 40 , the interconnection path  318 A( 1 ) electrically connects the inductor terminal  312 A( 1 ) to the device contact  316 A( 1 ) of the active semiconductor device  314 A( 1 ). The capacitive structure  310 A( 1 ) is formed in the interconnection path  318 A( 1 ) within the BEOL  330 . Similarly, with regard to the embodiment of the RF device  300 ( 2 ) shown in  FIG. 40 , the interconnection path  318 A( 2 ) electrically connects the inductor terminal  312 A( 2 ) to the device contact  316 A( 2 ) of the active semiconductor device  314 A( 2 ). The capacitive structure  310 A( 2 ) is formed in the interconnection path  318 A( 2 ) within the BEOL  330 . However, in this embodiment, the RF device  300 ( 1 ) and the RF device  300 ( 2 ) are interconnected using a trace  338  formed from the metallic layer M 4  within the BEOL  330 . Although the metallic layer M 4  is relatively thin, the trace  338  may be provided with the metallic layer M 1  when the RF device  300 ( 1 ) and the RF device  300 ( 2 ) are relatively close. If a distance between the RF devices  300 ( 1 ),  300 ( 2 ) is moderate, multiple metallic layers (thin and/or thick) can be stacked to reduce current path losses. 
       FIG. 41  illustrates an embodiment of the capacitive structure  310 , which may be provided as an embodiment of any of the capacitive structures  310 A,  310 B,  310 A( 1 ),  310 A( 2 ),  310 B( 1 ), or  310 B( 2 ) shown in  FIGS. 37-40 . The capacitive structure  310  is a metal-insulator-metal capacitive structure formed within an interconnection path  318 , which may be provided as an embodiment of any of the interconnection paths  318 A,  318 B,  318 A( 1 ),  318 A( 2 ),  318 B( 1 ), or  318 B( 2 ) shown in  FIGS. 37-40 . A top plate  340  of the capacitive structure  310  is formed by the metallic layer M 2  of the BEOL  330 , while a bottom plate  342  is formed by the metallic layer M 3  of the BEOL  330 . A dielectric layer DM is formed between the top plate  340  and the bottom plate  342  in the capacitive structure  310 . Since the metallic layer M 2  and the metallic layer M 3  are relatively thin, horizontal current flow would cause current crowding and result in Q factor degradation. Thus, the capacitive structure  310  provides a MIM configuration in which the current  313  flows vertically through the capacitive structure  310 . To prevent current circulation, an array of conductive vias  344  is formed on a bottom surface of the bottom plate  342  to split the current  313 . In this manner, the current  313  propagates through the interconnection path  318  to or from the semiconductor substrate  329  (shown in  FIG. 37 ). With the MIM capacitive structure, the current  313  propagates vertically in the metal and isolation material (change in polarization). 
       FIG. 42  illustrates another embodiment of the capacitive structure  310 , which may be provided as an embodiment of any of the capacitive structures  310 A,  310 B,  310 A( 1 ),  310 A( 2 ),  310 B( 1 ), and  310 B( 2 ), shown in  FIGS. 37-40 . The capacitive structure  310  is a metal-oxide-metal (MOM) capacitive structure formed within an interconnection path  318 , which may be provided as an embodiment of any of the interconnection paths  318 A,  318 B,  318 A( 1 ),  318 A( 2 ),  318 B( 1 ), or  318 B( 2 ) shown in  FIGS. 37-40 . The MOM capacitive structure shown in  FIG. 42  has a vertical configuration so that the current  313  flows vertically through metallic components in the MOM capacitive structure. In this regard, the capacitive structure  310  shown in  FIG. 42  includes a top plate  346  formed by the metallic layer M 1  of the BEOL  330 . The metallic layer M 1  is exposed externally while a bottom plate  348  is formed by the metallic layer M 3  of the BEOL  330 . The capacitive structure  310  includes an array of traces  350  between the top plate  346  and the bottom plate  348 . The array of traces  350  is provided by the metallic layer M 2  within the BEOL  330 . The traces in the array of traces  350  are electrically connected to the top plate  346  and the bottom plate  348  by conductive vias CVC. In this manner, the current  313  is divided when flowing through the conductive vias CVC and the array of traces  350 . 
     An oxide layer OM is formed between the top plate  346 , the bottom plate  348 , and the array of traces  350  in the capacitive structure  310 . Since the array of traces  350  is formed by the metallic layer M 2 , the traces  350  are substantially parallel to one another. Accordingly, the current flow of the current  313  is horizontal with respect to the oxide layer OM. In the oxide layer OM, there is no carrier conductor; rather, there is a polarization change. Since the current  313  propagates vertically through the top plate  346 , the conductive vias CVC, the array of traces  350 , and the bottom plate  348 , current crowding is avoided. While the current flow is horizontal through the oxide layer OM, this does not cause significant current crowding and does not significantly degrade a Q factor of the capacitive structure  310 . 
       FIG. 43  illustrates a top view of another embodiment of a capacitive structure  310 ( 1 ), which may be provided as an embodiment of any of the capacitive structures  310 A,  310 B,  310 A( 1 ),  310 A( 2 ),  310 B( 1 ), and  310 B( 2 ), shown in  FIGS. 37-40 ; a capacitive structure  310 ( 2 ); and a capacitive structure  310 ( 3 ). Each of the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) is a metal-to-metal (MTM) capacitive structure formed within an interconnection path  318 , which may be provided as an embodiment of any of the interconnection paths  318 A,  318 B,  318 A( 1 ),  318 A( 2 ),  318 B( 1 ), or  318 B( 2 ) shown in  FIGS. 37-40 . Thus, in this embodiment, the interconnection path  318  includes the multiple capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ). The MTM capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) can be formed using very few metallic layers, potentially resulting in low costs and a shorter interconnection path  318 . 
     In this embodiment, the interconnection path  318  is electrically connected and formed directly over active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ), which in this embodiment are FETs. More specifically, the interconnection path  318  is electrically connected to drain contacts D of each of the active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ). The active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ) also include gate contacts G and the source contacts S. The drain contacts D and the source contacts S are formed as interleaved metallic fingers. It should be noted that the MTM capacitive structure can be implemented on one or more of the device contacts D, S, G of each of the active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ). 
     Each of the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) shown in  FIG. 43  is formed using parallel metallic walls formed across the drains D of the active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ). Alternating metallic walls of the capacitive structure  310 ( 1 ) are electrically connected to the drain D of the active semiconductor device  314 ( 1 ) while the other metallic walls are electrically connected to a metallic layer (not shown) in the interconnection path  318  above the capacitive structure  310 ( 1 ). As a length of each of the metallic walls is increased, the capacitance of the capacitive structure  310 ( 1 ) is increased. With regard to the capacitive structure  310 ( 2 ), alternating metallic walls of the capacitive structure  310 ( 2 ) are electrically connected to the drain D of the active semiconductor device  314 ( 2 ) while the other metallic walls are electrically connected to a metallic layer (not shown) in the interconnection path  318  above the capacitive structure  310 ( 2 ). As each of a length of each of the metallic walls is increased, the capacitance of the capacitive structure  310 ( 2 ) is increased. Finally, alternating metallic walls of the capacitive structure  310 ( 3 ) are electrically connected to the drain D of the active semiconductor device  314 ( 3 ) while the other metallic walls are electrically connected to a metallic layer (not shown) in the interconnection path  318  above the capacitive structure  310 ( 3 ). As each of a length of each of the metallic walls is increased, the capacitance of the capacitive structure  310 ( 3 ) is increased. In this embodiment, each of the active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ) is configured to be turned on and off independently. As such, the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) can provide an overall digitally programmable capacitive structure, such as a digital array of capacitors. The MTM capacitive structure can also be realized using directly active or passive device interconnect metal layers requiring no extra metal layers or fewer metal layers. 
       FIG. 44  illustrates a top view of another embodiment of the capacitive structures  310 ( 1 ),  310 ( 2 ), and  310 ( 3 ). As in  FIG. 43 , each of the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) in  FIG. 44  is an MTM capacitive structure formed within the interconnection path  318 . In the embodiment shown in  FIG. 44 , the interconnection path  318  is electrically connected and formed directly over the same active semiconductor devices  314 ( 1 ),  314 ( 2 ), and  314 ( 3 ) described above with respect to  FIG. 43 . However, instead of the parallel metallic walls, each of the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) is formed using alternating posts encircled by metallic cages. The advantage of the capacitive structures  310 ( 1 ),  310 ( 2 ),  310 ( 3 ) shown in  FIG. 44  is that capacitances are realized in all directions and a capacitive density is determined by perimeters of the metallic cages. 
       FIG. 45  illustrates another embodiment of the capacitive structure  310 ( 1 ). As in  FIGS. 43 and 44 , the capacitive structure  310 ( 1 ) shown in  FIG. 45  is an MTM capacitive structure. The capacitive structure  310 ( 1 ) is formed using a metal interconnect. In this embodiment, the capacitive structure  310 ( 1 ) is formed using parallel metallic walls formed across the drain contact D and the source contact S of the active semiconductor device  314 ( 1 ). Alternating metallic walls of the capacitive structure  310 ( 1 ) are electrically connected to the drain contact D of the active semiconductor device  314 ( 1 ), while the other metallic walls are electrically connected to the source contact S of the active semiconductor device  314 ( 1 ). This embodiment is particularly advantageous in silicon-on-insulator (S 01 )-type processes and silicon-oxide-silicon (SOS)-type processes because drain-to-source diffusions are small. Furthermore, parasitic capacitances to the gate contact G of the active semiconductor device  314 ( 1 ) are less critical. As such, diffusions can be increased significantly without significantly degrading a Q factor of the capacitive structure  310 ( 1 ). If one side of the active semiconductor device  314 ( 1 ) is grounded, only the diffusion on that side may be increased. This provides an example of an asymmetric capacitive structure. 
       FIG. 46  illustrates yet another embodiment of the capacitive structure  310 ( 1 ). As in  FIGS. 43-45 , the capacitive structure  310 ( 1 ) shown in  FIG. 46  is an MTM capacitive structure. The capacitive structure  310 ( 1 ) is formed using a metal interconnect. The capacitive structure  310 ( 1 ) shown in  FIG. 46  is similar to the embodiment of the capacitive structure  310 ( 1 ) shown in  FIG. 45 . However, in this embodiment, the capacitive structure  310 ( 1 ) is also built using parallel metallic walls connected to the gate contact G in addition to the parallel metallic walls electrically connected to the source contact S and the drain contact D of the active semiconductor device  314 ( 1 ). This embodiment may be used with a planar semiconductor process, such as Complementary-Metal-Oxide Semiconductor (CMOS), Silicon Germanium (SiGe), Gallium Arsenide (GaAs), or Indium Phosphorus (InP)-type processes. The active semiconductor device  314 ( 1 ) is a FET. However, in alternative embodiments, the active semiconductor device  314 ( 1 ) may be a bipolar junction transistor (BJT), a heterojunction bipolar transistor (HBT) diode, or the like. 
     Those skilled in the art will recognize improvements and modifications to the preferred embodiments of the present disclosure. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.