Patent Publication Number: US-2009224685-A1

Title: Drive circuit for driving a gas discharge lamp, and method of calibrating a drive circuit

Description:
FIELD OF THE INVENTION 
     The invention relates in general to an electronic DC/AC drive circuit for driving an operational current in a load. The invention particularly relates to such a circuit for operating a lamp, specifically a gas discharge lamp, more specifically a high-pressure gas discharge lamp. The invention will hereinafter be explained in more detail with reference to a high-pressure gas discharge lamp, but this is by way of example only and should not be interpreted as limiting the scope of the invention. 
     BACKGROUND OF THE INVENTION 
     High-pressure gas discharge lamps should ideally be operated with an alternating current, so that, on a time scale larger than the period of the alternating current, the average DC level of the current is zero. Electronic circuits have been developed, which are capable of generating suitable lamp currents, in accordance with different designs. One category of such electronic circuits is designed to generate a commutating current, derived from a constant input voltage. 
     The invention specifically relates to an electronic lamp drive circuit of the type which comprises two independently controlled half-bridges, one half-bridge operating as a down-converter, and the other half-bridge operating as a commutator. Such a type of electronic lamp drive circuit will hereinafter be indicated as Combined Down-Converter Commutator Drive circuit, CDCCD circuit for short. 
     Examples of a CDCCD circuit are disclosed in WO-03/056886. Each half-bridge comprises two switches connected in series; the node between these switches constitutes an output of the corresponding bridge. A series arrangement of a first inductor, a lamp and a second inductor is connected between the two bridge output nodes. A controller controls the switches on the basis of a signal received from a current sensor, which senses the current through the first inductor. The controller also receives a reference signal. As a result of the switching action of the switches, the lamp current drops and rises at a relatively high frequency, so that the average lamp current follows the waveform of the reference signal. The reference signal, in turn, is generated in such a way that the average level of the lamp current is zero. 
     For a more elaborate explanation of the operation of the circuit, reference is made to WO-03/056886, the contents of which are herein incorporated by reference. 
     An important aspect of a correctly functioning CDCCD circuit is the accuracy of the current sensor, especially around the zero average lamp current. In practice, it may happen that a current sensor shows a small offset, which means that the output signal is not exactly zero when the measured current is actually equal to zero. Furthermore, current sensors are not exactly equal to each other, i.e. different current sensors may have different offsets. The controller has such a control action that the average measuring signal is zero. However, if the measuring signal is not proportional to the lamp current, especially if the measuring signal is offset with respect to the measured current, then the actual average current is not equal to zero. This situation would be very disadvantageous for the lamp driver as well as for the lamp, as it may increase power losses and shorten the maximum life of the driver and/or the lamp. 
     A further important aspect is that the sensor offset may change for any reason during operation, for instance, by thermal, mechanical, or magnetical influences, etc. Especially in the first minutes after lamp ignition, the largest thermal changes are expected. 
     A general objective of the invention is to improve the known CDCCD circuit and the accuracy of the current sensor. 
     SUMMARY OF THE INVENTION 
     In accordance with a first aspect of the invention, the controller is capable of operating in a calibration mode before the ignition mode. In the calibration mode, the zero level of the current sensor is detected. During the normal operational mode, the controller takes into account the offset characteristics of the sensor as determined during the calibration mode. 
     In a specific embodiment, the current reference signal for the controller is generated by a controllable reference signal generator, whose setting is controllable by the controller. The CDCCD circuit further comprises a voltage sensor, measuring the lamp voltage. In the calibration mode, the controller drives the switches in such a way that an alternating lamp voltage is generated, while ensuring that no lamp current flows. The controller adjusts the setting of the reference signal generator in such a way that the average output voltage is equal to half the value of the input voltage. During the normal operational mode, the reference signal generator operates with the adjusted setting. 
     In a preferred embodiment, the controller keeps the switches of the commutating half-bridge in their OFF state during the calibration mode in order to ensure that no current can flow through the lamp. 
     In accordance with a second aspect of the invention, the controller is capable of operating in a recalibration mode during the normal operational mode. In the recalibration mode, the normal operation is briefly interrupted, so that the lamp current is zero, and a calibration measurement is performed, after which normal operation is resumed. The interruption is much shorter than half the current period, so that the lamp immediately ignites when normal operation is resumed, and the brief interruption of the light is hardly noticeable to the human eye. The recalibration mode is performed during positive current periods as well as during negative periods, and the results are combined to calculate an adjusted setting for the reference signal generator. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       These and other aspects, features and advantages of the invention will be further explained by means of the following description with reference to the drawings, in which identical reference numerals indicate identical or similar parts, and in which: 
         FIG. 1  is a block diagram showing a CDCCD circuit according to the invention; 
         FIG. 2  is a graph showing the lamp current as a function of time; 
         FIG. 3  is a graph showing the lamp current as a function of time on a larger time scale; 
         FIG. 4A  is a graph illustrating an offset of a current sensor; 
         FIG. 4B  is a graph illustrating a consequence of a current sensor offset; 
         FIG. 5  is a graph illustrating an effect of a shifted reference signal; 
         FIGS. 6A-B  are block diagrams illustrating alternative embodiments of a CDCCD circuit according to the invention; 
         FIG. 7  is a graph illustrating the AC lamp current and the current-measuring signal during a calibration mode according to the invention; 
         FIG. 8  is a graph showing a voltage-measuring signal as a function of time; 
         FIG. 9  is a graph showing the current during a recalibration sequence. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  is a block diagram showing a CDCCD circuit  100  according to the invention. The CDCCD circuit  100  has a first input terminal  101  and a second input terminal  102  for connection to an input voltage source (not shown), which is expected to provide a DC voltage VDC wherein the first terminal  101  is positive with respect to the second terminal  102 . 
     The CDCCD circuit  100  comprises a first switching bridge  110  and a second switching bridge  120 , connected in parallel between said first and second input terminals  101 ,  102 . The first bridge  110  comprises a series arrangement of a first controllable switch  111  and a second controllable switch  112 , the node  113  between these two switches  111 ,  112  constituting a bridge output node. Likewise, the second bridge  120  comprises a series arrangement of a third controllable switch  121  and a fourth controllable switch  122 , the node  123  between these two switches constituting an output node of the second bridge. As illustrated, the controllable switches are suitably implemented as MOSFETS. 
     The CDCCD circuit  100  has a first load output terminal  191  and a second load output terminal  192  for connecting a load L. In the illustration of  FIG. 1 , a lamp L is connected between these two output terminals  191 ,  192 . In the following description, the operation of the CDCCD circuit  100  will be further explained with reference to a lamp as a load, but it should be recognized that the CDCCD circuit  100  can be used for driving other types of loads. 
     The CDCCD circuit  100  further comprises a first inductor  131 , for instance, a coil, connected between the first bridge output node  113  and the first load output terminal  191 , and a second inductor  132 , for instance, a coil, connected between the second bridge output node  123  and the second load output terminal  192 . Furthermore, the CDCCD circuit  100  comprises a first capacitor  141  connected between the first load output terminal  191  and the second input terminal  102 , and a second capacitor  142  connected between the second load output terminal  192  and the second input terminal  102 . Alternatively, one or both of the first and second capacitors  141 ,  142  may be connected to the first input terminal  101 , or to any other source of constant potential. 
     The CDCCD circuit  100  further comprises a current sensor  150  arranged to measure the current in the first inductor  131 , and designed to generate a current-measuring signal S 1  representing the measured current. In the embodiment as illustrated, the current sensor  150  is shown at a position associated with a current-conducting line  151  connecting the first inductor  131  with the first load output terminal  191 , thus actually measuring the current between the inductor  131  and the output terminal  191 . However, it should be noted that this current is identical to the current in the inductor  131 . Furthermore, it should be noted that alternative locations of the current sensor  150  are also possible. 
     The measuring signal S 1  is received at a sensor input  176  of a switch controller  170 , which also has a reference input  177  receiving a current reference signal SR generated by a current reference signal generator  160 . The switch controller  170  has four control outputs  171 ,  172 ,  173 ,  174 , coupled to control inputs of the controllable switches  111 ,  112 ,  121 ,  122 , respectively. The switch controller  170  is designed to generate control signals SC 1 , SC 2 , SC 3 , SC 4  for the four controllable switches  111 ,  112 ,  121 ,  122 , respectively, in order to control the operative state of these four switches on the basis of the current reference signal SR and the current-measuring signal S 1 , as will be explained in more detail below. 
     Each controllable switch has two operative states: a first operative state in which the switch is conductive, and a second operative state in which the switch is non-conductive. In the following description, the conductive state of a switch will also be indicated as ON or CLOSED, whereas the non-conductive state of a switch will be indicated as OFF or OPEN. 
     Furthermore, a control signal resulting in a switch being open or closed, respectively, will also be indicated as an OPEN signal or a CLOSED signal, respectively. 
     In normal operation, as will be explained in more detail, the switches of a bridge are controlled to have mutually opposite operative states. This wording is used to indicate that one switch is OPEN, whereas the other is CLOSED, and vice versa. It follows that the bridge as a whole has a first bridge-operative state wherein the switch connecting the output node to the high voltage input terminal  101  is ON, whereas the other switch is OFF, and a second bridge-operative state wherein the switch connecting the output node to the low voltage input terminal  102  is ON, whereas the other switch is OFF. These two bridge-operative states will be indicated as the HIGH state and the LOW state, respectively. 
     The switching bridges  110 ,  120  actually also have a third operative state wherein both switches are ON, and a fourth operative state wherein both switches are OFF. A person skilled in the art will recognize that the third operative state, which will be indicated as the SHORT state, is to be avoided because it constitutes a short circuit between the high voltage input terminal  101  and the low voltage input terminal  102 . Therefore, the switch controller  170  is designed to generate its control signals for the two switches of one bridge, so that, at a transition from a HIGH bridge state to a LOW bridge state or vice versa, the ON switch is first opened while the OFF switch is closed with a brief delay, so that the transition takes place via the fourth operative state, which will be indicated as the OFF state. 
     As explained more elaborately in WO-03/056886, the switch controller  170  is capable of operating in three different modes for operating a high-pressure gas discharge lamp, i.e. an ignition mode, a run-up mode, and a normal operational mode. For an explanation of these modes, reference is made to said publication. As far as is relevant for the invention, the operation of the switch controller  170  is explained in more detail with reference to the normal operational mode. 
       FIG. 2  is a graph showing the lamp current (vertical axis) as a function of time (horizontal axis). The fourth switch  122  is assumed to be in the ON state. 
     In the normal operational mode, the first bridge  110  is switched from its HIGH bridge state to its LOW bridge state at a relatively high frequency, typically of the order of about 300 kHz. 
     The lamp current through the lamp L flows in the direction from the first bridge  110  to the second bridge  120 . At instant t 1 , the first bridge  110  is switched to its HIGH state, and the lamp current increases from a low value I 1  to a higher value I 2  at instant t 2 , when the first bridge  110  is switched back to its LOW state. From instants t 2  to t 3 , the lamp current decreases from the high value I 2  to the low value I 1 . The above process is repeated as from instant t 3 . On a time scale larger than (t 3 -t 1 ), the lamp current has an average value Iav, indicated in  FIG. 2  as a horizontal line. The level of this average lamp current Iav is controlled by the switch controller  170  by suitably setting the duty cycle of the first bridge  110 , i.e. the ratio of (t 2 -t 1 ) to (t 3 -t 1 ). 
     The above process continues until the second bridge  120  is switched from its LOW state to its HIGH state. Again, the lamp current increases and decreases at a frequency determined by the switching frequency of the first bridge  110 , which is also indicated as the down-converter bridge, but now the direction of the lamp current is reversed, so that the lamp current flows from the second bridge  120  to the first bridge  110 .  FIG. 3  is a graph comparable to  FIG. 2 , but now on a larger time scale, showing how the average lamp current Iav (vertical axis) changes direction at a frequency determined by the switching frequency of the second bridge  120 , also indicated as commutator bridge. More specifically,  FIG. 3  illustrates that, before instant t 6 , when the commutator bridge  120  is in its LOW state (the situation in  FIG. 2 ), the average lamp current Iav has a first direction, arbitrarily indicated as positive direction, and a first magnitude indicated as I P , while after instant t 6 , when the commutator bridge  120  is in its HIGH state, the average lamp current has the opposite direction, indicated as negative direction, and a second magnitude indicated as I N . This situation continues until instant t 7 , when the commutator bridge  120  switches back to its LOW state and the average lamp current Iav switches back to the positive direction and magnitude I P . This process is repeated with a commutating frequency determined by the switching frequency of the commutator bridge  120 , which typically is of the order of about 100 Hz. 
     The switch controller  170  generates its control signals SC 1 , SC 2 , SC 3 , SC 4  for the four switches  111 ,  112 ,  121 ,  122  on the basis of its input signals received at its inputs  176  and  177 . The current reference signal generator  160  generates the current reference signal S R , so that it represents the desired waveform of the lamp current. Typically, this desired waveform is a square wave with a 50% duty cycle and a zero DC level. The control signals for the switches are generated in such a way that the current-measuring signal S 1  provided by the current sensor  150  follows this current reference signal S R . In  FIG. 3 , the current reference signal S R  is also shown. It can be seen in  FIG. 3  that the current reference signal S R  is a customarily symmetrical signal having a 50% duty cycle and a zero DC level, corresponding to the desired waveform of the lamp current. 
     Ideally, the current sensor  150  has a linear characteristic, indicated by the dotted line  41  in  FIG. 4A , which shows a graph of sensor output signal S 1  (vertical axis) versus actual measured current I (horizontal axis). However, in practice, it may happen that the current sensor  150  shows an offset Δ, such that its characteristic is represented by line  42  in  FIG. 4A : if the current is equal to zero, the sensor output signal S 1  has a value Δ, and the sensor output signal S 1  is equal to zero only when the actual current has a magnitude I A . This constitutes a problem, as is illustrated in  FIG. 4B . If the current reference signal S R  was a symmetrical signal having a 50% duty cycle and a zero DC level, and if the switch controller  170  was operated in such a way that the sensor output signal S 1  is made to follow the reference signal S R , the lamp current would have a DC level equal to I A , i.e. unequal to zero. It is to be noted that, in this case, the sensor output signal S 1  would have a value A, so the switch controller  170  would believe that the operation is OK, but the sensor output signal does not accurately represent the actual current, which suffers from a DC offset. 
     According to the invention, the control action of the switch controller  170  is manipulated in such a way that the actual current has the desired waveform of a 50% duty cycle and a zero DC level while the sensor output signal S 1  does not have this desired waveform. In accordance with a first aspect of the invention, the reference signal S R  is shifted over a distance Δ C , to obtain a shifted reference signal S R ′=S R +Δ C  as illustrated in  FIG. 5 , and the operation of the switch controller  170  is such that the sensor output signal S 1  is made to follow the shifted reference signal S R ′. In such a case, of course, the sensor output signal S 1  now has a DC level Δ which is offset with respect to zero, corresponding to the offset Δ C  of the reference signal S R . However, the average lamp current Iav now has a DC level which is substantially equal to zero. 
     In the embodiment illustrated in  FIG. 1 , the current reference signal generator  160  is a controllable signal generator having a control input  161  coupled to a fifth control output  175  of the switch controller  170 , and the switch controller  170  is designed to generate a reference control signal SC R  for the signal generator  160  at its fifth output  175 . The signal generator  160  is adapted to generate its reference signal S R  with an offset Δ C  as determined by the reference control signal SC R  received at its control input  161 . 
       FIG. 6A  is a block diagram which is comparable to  FIG. 1  and illustrates an alternative embodiment, in which the signal generator  160  does not need to be a controllable generator: in this case, the signal generator  160  is designed to generate a symmetrical current reference signal S R  as usual. For the sake of simplicity, only the switch controller  170  and the signal generator  160  are shown in  FIG. 6A . The switch controller  170  is provided with an adder  180  having a first input  186  receiving the current reference signal S R  from the signal generator  160 . The switch controller  170  has an offset output  178  providing an offset signal Δ C , which is received by the adder  180  at a second input  188 . The adder  180  adds the two signals received at its two inputs  186  and  188 , and generates at an output  187  a corrected current reference signal S R ′ which is equal to the summation of the original reference signal S R  from the reference signal generator  160  and the offset signal Δ C  provided by the switch controller  170 , which output  187  is coupled to the reference input  177  of the switch controller  170 . 
     In a modification, the adder  180  is an integral part of the switch controller  170 . 
     In another approach of the invention, the sensor output signal S 1  is shifted over a distance Δ in order to compensate the offset in this signal. An embodiment implementing this approach is illustrated in  FIG. 6B . The switch controller  170  is provided with a subtractor  190  having a first input  198  receiving the sensor output signal S 1   R  from the sensor  150 . The switch controller  170  has an offset output  179  providing an offset signal A, which is received by the subtractor  190  at a second input  199 . The subtractor  190  is designed to subtract the signal received at its second input  199  from the signal received at its first input  198 , and generates at an output  196  a corrected current sensor signal S 1 ′=S 1 −Δ which is equal to the difference between the original sensor output signal S 1  from the current sensor  150  and the offset signal Δ provided by the switch controller  170 , which output  196  is coupled to the sensor input  176  of the switch controller  170 . 
     In a modification, the subtractor  190  is an integral part of the switch controller  170 . 
     In order to be able to determine a suitable value for the control signal S CR  (embodiment of  FIG. 1 ), or for the reference signal offset Δ C  (embodiment of  FIG. 6A ), or for the sensor correction signal Δ (embodiment of  FIG. 6B ), the switch controller  170  is capable of operating in a calibration mode, as will be explained in the following description. In the calibration mode, the switch controller  170  is set to generate a symmetrical lamp voltage in the absence of a lamp current. As a result, if the same setting is used to generate a lamp current, the average lamp current will be zero. 
     The switch controller  170  executes the calibration mode before the ignition mode, so the lamp L has not ignited yet, and no current can flow through the lamp L. However, in practice, it may happen that some spurious current flows erratically through the lamp L. Furthermore, as mentioned above, the invention is also applicable to cases where the load L is not a discharge lamp, so in general it may happen that the load L is conductive even before the ignition mode. Therefore, in order to prevent any current from flowing through the load L, the switch controller  170  is preferably designed to switch the commutator bridge  120  to its OFF state during the calibration mode. 
     Thus, it is ensured that no current can flow through the first inductor  131 , because such a current would have to flow either through the load L (which is inhibited as explained above) or through the first capacitor  141  (which is inhibited by the characteristics of the first capacitor  141 ). 
     In the calibration mode, the switch controller  170  switches the down-converter bridge  110  from its HIGH state to its LOW state at a relatively high frequency, typically equal to the operation frequency of the down-converter bridge  110  during the normal operational mode. As a result, an AC current I L  is generated in the current path from the first bridge output  113  via the first inductor  131  and the first capacitor  141 , which is an AC current without any DC component. Thus, as illustrated in  FIG. 7 , the sensor output signal S 1  should now be representative of an AC current without a DC component: any DC component of the current sensor output signal S 1  is due to an offset of the current sensor  150 , i.e. is equal to the offset Δ in  FIG. 4A . Thus, the switch controller  170  is capable of actually measuring the current sensor offset Δ. 
     For compensating the current sensor  150 , the invention uses the voltage at the first output terminal  191 . To this end, as illustrated in  FIG. 1 , the CDCCD circuit  100  comprises a voltage sensor  155  having a sense input  156  connected to the first output terminal  191 , and a signal output  157  coupled to a signal input  158  of the switch controller  170 . By way of example, the voltage sensor  155  may be implemented as a resistance divider. 
       FIG. 8  is a graph showing the voltage-measuring signal S 2  as a function of time (curve  81 ).  FIG. 8  also shows the voltage level V 101  at the first input terminal  101  (horizontal line  82 ), and the voltage level V 102  at the second input terminal  102  (horizontal line  83 ). These voltage levels V 101  and V 102  are also received by the switch controller  170 , but this is not shown in the drawings. 
     The voltage-measuring signal S 2  is shown as a square-wave signal  81  having a top level V T  which is lower than the first input voltage level V 101 , and a minimum value V L  which is higher than the second input voltage level V 102 . This is, however, not essential. 
     During a HIGH state of the down-converter bridge  110 , the switch controller  170  measures the difference between the voltage-measuring signal S 2  and the first input voltage level V 101 . The absolute value of the result of this measurement is indicated in  FIG. 8  as voltage difference V A . 
     During the subsequent LOW state of the down-converter bridge  110 , the switch controller  170  measures the difference between the voltage-measuring signal S 2  and the second input voltage level V 102 . The absolute value of the result of this measurement is indicated in  FIG. 8  as V B . Ideally, the lamp voltage at the first output terminal  191  should be symmetrical with respect to the input voltage levels V 101  and V 102 . This means that V A  should be equal to V B . If V A  is not equal to V B , a correction is required so as to reduce the difference V A −V B . 
     In the embodiment of  FIG. 1 , the switch controller  170  generates its reference control signal SC R  for the current reference signal generator  160  in such a way that the reference signal outputted by the current reference signal generator  160  is shifted (S R (ΔC); see  FIG. 5 , top graph), shifting the voltage at the first output terminal  191  so as to reduce the difference V A −V B . 
     The above steps are then repeated until said difference V A −V B  is equal to zero within a certain predefined range of tolerances. 
     The value of the reference control signal SC R  thus obtained is maintained by the switch controller  170  in the subsequent ignition, run-up, and normal operational modes. 
     In the embodiment of  FIG. 6A , the switch controller  170  generates its offset signal Δ C  for the adder  180  in such a way that the corrected reference signal S R ′ outputted by the adder  180  is shifted with respect to the original reference signal S R  from the reference signal generator  160  (S R ′=S R +ΔC); see  FIG. 5  (top graph), shifting the voltage at the first output terminal  191  so as to reduce the difference V A −V B . 
     The above steps are then repeated until said difference V A −V B  is equal to zero within a certain predefined range of tolerances. 
     The value of the offset signal Δ C  thus obtained is maintained by the switch controller  170  in the subsequent ignition, run-up, and normal operational modes. 
     In the embodiment of  FIG. 6B , the switch controller  170  generates its offset signal Δ for the subtractor  190  in such a way that the signal S 1 ′ received at its sensor input  176  is equal to zero within a certain predefined range of tolerances. 
     The value of the offset signal Δ thus obtained is maintained by the switch controller  170  in the subsequent ignition, run-up, and normal operational modes. 
     During normal operation, it may happen that the offset of the current sensor changes; especially in the first minutes after lamp ignition, the temperature of the driver is expected to change and, as a result, offset changes of the current sensor are expected. It is noted that it is not possible for the driver to switch to the calibration mode as described above, because then the lamp would extinguish. 
     In accordance with a further aspect of the invention, the switch controller  170  is capable of operating in a recalibration mode during the normal operational mode. In this recalibration mode, the switch controller  170  alternates normal operation with calibration measurement operation, as illustrated in  FIG. 9 .  FIG. 9  is a graph showing the load current I L  as a function of time, on a time scale comparable to the time scale of  FIG. 3 . At instant t 10 , when the switch controller  170  is in its normal operation, the commutator bridge  120  is switched to its LOW state (compare instant t 7  in  FIG. 3 ). The subsequent commutation instants are instants t 20  and t 30 . The phase from instant t 10  to instant t 20  will be indicated as the positive current period, whereas the phase from instant t 20  to instant t 30  will be indicated as the negative current period; the phase from t 10  to t 30  will be indicated as the entire current period. 
     At instant t 11  during the positive current period, the switch controller  170  enters a calibration measurement operation by switching the down-converter bridge  110  to its OFF state. Instant t 11  is preferably chosen to be such that (t 11 -t 10 ) is approximately equal to 10%-30% of (t 20 -t 10 ). 
     The energy in the system discharges via the commutator bridge  120 , which takes about 100 to 200 μsec, depending on the actual circuit design, as should be clear to a person skilled in the art. Then, no DC current can flow in the load L any more. To make sure that no current can flow in the load L, indeed, the commutator bridge  120  is switched to its OFF state at instant t 12 . Then, starting at t 13 , the down-converter bridge  110  is operated again at a high frequency, preferably the same frequency as during normal operation, producing an AC current in the first inductor  131  and the first capacitor  141 , which AC current has a zero DC level. 
     At instant t 14 , the commutator bridge  120  is switched to its LOW state again, so as to end the calibration measurement operation and to resume normal operation. The duration from instant t 13  to instant t 14 , which will be indicated as the AC current phase of the calibration measurement operation, may typically be of the order of about 100 μsec. 
     During the calibration measurement operation, the lamp L is off. The entire calibration measurement operation from instant t 11  to instant t 14  has a very short duration, typically less than 500 μsec, so that, at instant t 14 , the lamp L is still hot enough to re-ignite immediately. Furthermore, the normal lamp operation is interrupted so briefly that it is not disturbing to the human eye. In any case, the calibration measurement operation from instant t 11  to instant t 14  falls entirely within the positive current period. 
     During the AC current phase of the calibration measurement operation, the switch controller  170  receives the current-measuring signal S 1  from the current sensor  150 , and calculates the DC level of the current-measuring signal S 1 . This DC level during the positive current period will be indicated as DC[+]. 
     In a similar manner, a calibration measurement operation is performed from instant t 21  to instant t 24  during a negative current period. Again, the DC level of the current-measuring signal S 1  is calculated; this DC level during the negative current period will be indicated as DC[−]. Although it is possible that one or more “uninterrupted” current periods are passing between these two calibration measurement operations, it is preferred that this subsequent calibration measurement operation is performed in the negative current period which immediately follows the positive current period t 10 -t 20 , as illustrated. 
     The above-described sequence of a calibration measurement operation during a positive current period and a calibration measurement operation during a subsequent negative current period will be indicated as a calibration measurement sequence. As already mentioned, a calibration measurement sequence preferably takes place during one full current period. 
     Although, in principle, it may be sufficient to have only one calibration measurement sequence, it is preferred to repeat the calibration measurement sequence a few times, for instance, 10 times. The combination of these calibration measurement sequences will be indicated as a calibration measurement cycle. The calibration measurement sequences may be performed in subsequent full current periods, but it is also possible that one or more positive or negative current periods are skipped before the next calibration measurement sequence. 
     Each calibration measurement sequence will yield a value for DC[+] and a value for DC[−]. Thus, the calibration measurement cycle will yield a plurality of values for DC[+]; the average of these values will be indicated as &lt;DC[+]&gt;. Likewise, the calibration measurement cycle will yield a plurality of values for DC[−]; the average of these values will be indicated as &lt;DC[−]&gt;. 
     If the current sensor  150  operates free from any offset, said average values &lt;DC[+]&gt; and &lt;DC[−]&gt; will be equal to zero. A sensor calibration correction value SCC will be calculated as SCC=α·(&lt;DC[+]&gt;+&lt;DC[−]&gt;)/2, wherein α is a factor which may be predetermined, or determined empirically. 
     In the next step, the switch controller  170  will adjust the current sensor correction setting, using said sensor calibration correction value SCC. 
     For instance, in the embodiment of  FIG. 1 , the switch controller  170  will adjust the reference control signal SC R  for the current reference signal generator  160  in accordance with 
       SC R (NEW)=SC R (OLD)+SCC. 
     In the embodiment of  FIG. 6A , the switch controller  170  will adjust the offset signal Δ C  for the adder  180  in accordance with 
       Δ C (NEW)=Δ C (OLD)+SCC. 
     In the embodiment of  FIG. 6B , the switch controller  170  will adjust the offset signal Δ for the subtractor  190  in accordance with 
       Δ(NEW)=Δ(OLD)+SCC. 
     It should be clear that the offset of the current sensor  150  is not fully compensated if α is too small, whereas the offset of the current sensor  150  is overcompensated if α is too high. It is not necessary that α is exactly correct, as long as it is ensured that the offset after adjustment is smaller than before. Then, the offset can be reduced in subsequent steps by repeating the calibration measurement cycle a few times. The switch controller  170  may decide to quit the recalibration mode when it finds SCC to be smaller than a predetermined threshold. 
     The entire recalibration mode may last a relatively short time. If a calibration measurement cycle takes ten subsequent calibration measurement sequences, and if the calibration measurement cycle is performed ten times, the entire recalibration mode will take about one second, assuming that the commutation frequency is 100 Hz. 
     The recalibration mode is preferably performed repeatedly, wherein the intervals between subsequent recalibration modes may be relatively short (about 10 seconds to 1 minute) shortly after ignition, while the intervals between subsequent recalibration modes may increase later. Eventually, once the lamp has been burning for a sufficiently long time, it may be decided that the recalibration mode is no longer necessary. 
     It is also possible that means are provided for generating a signal which is indicative of a parameter of the environment, for instance, temperature. In such a case, such a parameter may be monitored, and a recalibration mode may be performed when such a parameter has changed by a certain predefined amount or a certain predefined percentage. 
     It should be clear to a person skilled in the art that the invention is not limited to the embodiments described above by way of example, and that several variations and modifications are possible within the protective scope of the invention as defined in the appendent claims. 
     The invention has been explained hereinbefore with reference to block diagrams, which illustrate functional blocks of the device according to the invention. It is to be understood that one or more of these functional blocks may be implemented in hardware, wherein the function of such a functional block is performed by individual hardware components, but one or more of these functional blocks may alternatively be implemented in software, so that the function of such a functional block is performed by one or more program lines of a computer program or a programmable device such as a microprocessor, a microcontroller, a digital signal processor, etc.