Patent Publication Number: US-6982590-B2

Title: Bias current generating circuit, laser diode driving circuit, and optical communication transmitter

Description:
CROSS REFERENCE TO RELATED APPLICATION 
   This application is based upon and claims benefit of priority under 35 USC 119 from the Japanese Patent Application No. 2003-124034, filed on Apr. 28, 2003, the entire contents of which are incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   The present invention relates to a bias current generating circuit, laser diode driving circuit, and optical communication transmitter. 
   A circuit for driving a laser diode amplifies a high-speed digital signal output from a time multiplexing circuit called a serializer or multiplexer, and outputs a driving current necessary to drive the laser diode. 
   This laser diode driving circuit is required not only to amplify a high-speed signal but also to supply a temperature-dependent driving current. 
   Generally, when the temperature rises, a laser diode increases an emission threshold current and its emission efficiency lowers. The emission threshold current is the value of a driving current for starting light emission. The emission efficiency is the value obtained by differentiating the optical output signal power amplitude by the driving current. 
   The increase in emission threshold current is suppressed by controlling the current value of a bias current generating circuit installed separately from a high-speed-signal processing amplifier in the laser diode driving circuit. This control is to receive an output current from a monitoring photodiode formed close to the laser diode, and adjust the bias current in accordance with the current value. 
   A direct current generated by this bias current generator determines the average emission power of the laser diode. The monitoring photodiode senses this average emission power, and the signal is returned to the bias current generator. By this negative feedback path, the average emission power can be controlled independently of the temperature. 
   A method of compensating for the decrease in emission efficiency caused by the temperature rise of the laser diode will be explained below. 
     FIG. 5  shows the arrangement of a high-speed-signal amplifier of the laser diode driving circuit. This high-speed-signal amplifier has a driving current controller  1 , differential output unit  2 , and differential amplifier DA 100  as a driving stage. 
   A pair of differential signals are input to a non-inverting input terminal IN+ and inverting input terminal IN− of the differential amplifier DA 100 . Through this driver stage, the signals are input to the differential output unit  2  as a final amplification stage and output from it. 
   The differential output unit  2  is a differential circuit ECL (Emitter Coupled Logic) including bipolar transistors Q 200  and Q 201 . However, the differential output unit  2  can also be constructed using FETs such as MESFETs or MOSFETs, instead of bipolar transistors. 
   The differential output unit  2  includes resistors R 100  and R 101 , the bipolar transistors Q 200  and Q 201  making a differential pair, and a bipolar transistor Q 202  as a current source transistor. 
   A bias current to be supplied to the base of the bipolar transistor Q 202  is controlled by the driving current controller  1 . Although the emitter of the bipolar transistor Q 202  is directly grounded in  FIG. 5 , this emitter may also be grounded via a resistor. 
   The driving current controller  1  has a bias-current generating circuit BGC 1  for generating a bias current Ibias, and bipolar transistors Q 100  and Q 101 , and forms a current mirror circuit together with the transistor Q 202  of the differential output unit  2 . 
   The bias current Ibias generated by the bias current generating circuit BGC 1  must be preset so as to rise at a desired ratio when the temperature rises, in order to meet the characteristics of the laser diode. 
   The conventional bias current generating circuit will be described below with reference to  FIG. 6 . 
   This bias current generating circuit comprises a bandgap reference circuit BGRC, low-potential-side, constant-current source circuits LCS 1  and LCS 2 , and a current mirror circuit. The bandgap reference circuit BGRC includes resistors R 1 , R 2 , R 3 , and R 4 , NPN transistors Q 1  and Q 2 , an N-channel transistor N 1 , and an operational amplifier OP 1 . The low-potential-side, constant-current source circuit LCS 1  includes an N-channel transistor N 3 , operational amplifier OP 4 , external terminal PAD 1 , and external resistor R 7 . The low-potential-side, constant-current source circuit LCS 2  includes an N-channel transistor N 4 , operational amplifier OP 5 , external terminal PAD 2 , and external resistor R 9 . The current mirror circuit includes P-channel transistors P 2  and P 3 . 
   The parameters of the resistors R 1 , R 2 , R 3 , and R 4 , NPN transistors Q 1  and Q 2 , N-channel transistor N 1 , and operational amplifier OP 1  are so set that the circuit including these elements operates as the bandgap reference circuit BGRC. 
   Accordingly, an output potential V 2  from the operational amplifier OP 1  maintains about 1.2 V independently of the temperature and a power supply voltage Vcc. In contrast to the potential V 2 , a contact potential V 1  proportional to absolute temperature is generated from the connection node between the resistors R 3  and R 4 . At room temperature, the potential V 1  is half (about 0.6 V) the potential V 2 . 
   The NPN transistor N 1  forms a startup circuit controlled by an activation signal Startup which momentarily changes to high level when the power supply is turned on and then rapidly goes to a ground potential Vss. The NPN transistor N 1  allows the bandgap reference circuit BGRC to reach a desired operating point immediately after the power supply is turned on. 
   Two constant-current source circuits which use the two potentials V 1  and V 2  generated by the bandgap reference circuit BGRC as reference potentials generate electric currents I 1  and I 2 , respectively. 
   That is, a first constant-current source circuit including the operational amplifier OP 4 , NPN transistor N 3 , and resistor R 7  generates the electric current I 1  (=V 1 /R 7 ), and a second constant-current source circuit including the operational amplifier OP 5 , NPN transistor N 4 , and resistor R 9  generates the electric current I 2  (=V 2 /R 9 ). The resistor R 7  is connected between the external terminal PAD 1  and ground voltage Vss, and the resistor R 9  is connected between the external terminal PAD 2  and ground voltage Vss. The resistors R 7  and R 9  are formed outside a semiconductor integrated circuit forming the laser diode driving circuit, and implemented by fixed resistors, variable resistors, electronic volume ICs, or the like. 
   The electric currents I 1  and I 2  are added to form an electric current  13  which functions as a reference current of the current mirror circuit formed by the two PMOS transistors P 2  and P 3 . As a consequence, the bias current Ibias amplified by the gate width ratio (M) of the PMOS transistor P 3  to the PMOS transistor P 2  is output as a mirror current. This bias current Ibias is the bias current Ibias finally output from the bias current generating circuit BGC 1  in the driving current controller  1  shown in  FIG. 5 . The transistors Q 100 , Q 101 , and Q 202  form a current mirror. The collector current of the transistor Q 202  of the differential output unit  2  is the value obtained by multiplying the size ratio of Q 202  to Q 101  by the reference current IIbias. Consequently, the laser diode driving current amplitude is proportional to the reference current Ibias. 
   From the foregoing, letting T denote absolute temperature, Ibias is represented by 
                   Ibias   =     M   ×   I   ⁢           ⁢   3                 =     M   ×     (       I   ⁢           ⁢   1     +     I   ⁢           ⁢   2       )                   =     M   ×     {       (     V   ⁢           ⁢     1   /   R     ⁢           ⁢   7     )     +     (     V   ⁢           ⁢     2   /   R     ⁢           ⁢   9     )       }                   =     M   ×     {       A   ×   T     +   B     }                     (   1   )             
 
where A and B are constants and represented by
 
 A ≅(0.002 /R   7 )× T   (2)
 
 B≅ 1.2 /R   9   (3)
 
     FIG. 7  shows an example of the temperature dependence of each of the electric currents I 1 , I 2 , and I 3 . 
   The ratio of the electric current I 1  to the electric current I 2  can be changed by the values of the resistors R 7  and R 9 . When the ratio of the electric current  12  is raised, the temperature dependence of the bias current Ibias decreases. When the ratio of the electric current I 1  is raised, the temperature dependence of the bias current Ibias increases. 
   As described above, by adjusting the values of the external resistors R 7  and R 9  in accordance with the temperature dependence of the emission efficiency of each individual laser diode, the optical output amplitude of the laser diode can be held constant regardless of the temperature. 
   The bias current Ibias of the bias current generating circuit shown in  FIG. 6  becomes zero at absolute zero, when the resistor R 9  is made infinite, i.e., when the resistor R 9  is removed. That is, this bias current generating circuit has characteristics proportional to the temperature. 
   If the bias current Ibias at a certain temperature To is regarded as a reference, the rate of increase of the bias current Ibias per degree of the temperature is 1/To. If the temperature To is room temperature (300K), the rate of change of the bias current Ibias to the temperature is 1/300≅3333PPM. 
   Generally, the temperature dependence of the emission efficiency of a laser diode is larger than 3333PPM. The laser diode driving circuit having the bias current generating circuit shown in  FIG. 6  cannot perform temperature compensation for such a laser diode. Accordingly, no optical signal power amplitude independent of the temperature can be obtained. 
   The following is a reference disclosing the conventional current control technique. 
   Japanese Patent Laid-Open No. 2000-244250. 
   As described above, the conventional bias current generating circuit cannot well perform temperature compensation for the temperature dependence of the emission efficiency of a laser diode. 
   SUMMARY OF THE INVENTION 
   According to one aspect of the present invention, there is provided a bias current generating circuit comprising, 
   a bandgap reference circuit connected to a high power supply voltage terminal for receiving a high power supply voltage and a low power supply voltage terminal for receiving a low power supply voltage, and having a first output terminal for outputting a first voltage which is constant regardless of a temperature, and a second output terminal for outputting a second voltage which changes in accordance with a temperature; 
   a first low-potential-side constant-current source circuit which includes a first resistor connected between said low power supply voltage terminal and a first terminal, and a first current path connected between said first terminal and a first current supply terminal, receives the second voltage as a reference potential, and outputs a first electric current dependent on a temperature and corresponding to said first resistor from said first current supply terminal; 
   a second low-potential-side constant-current source circuit which includes a second resistor connected between said low power supply voltage terminal and a second terminal, and a second current path connected between said second terminal and a second current supply terminal, receives the first voltage as a reference potential, and outputs a second electric current independent of a temperature and corresponding to said second resistor from said second current supply terminal; 
   a third resistor having one end connected to said high power supply voltage terminal; 
   a third low-potential-side constant-current source circuit which is connected between the other end of said third resistor and said low power supply voltage terminal, receives the first voltage as a reference potential, and supplies a temperature-independent third electric current to said third resistor; 
   a high-potential-side constant-current source circuit which includes a fourth resistor connected between said high power supply voltage terminal and a third terminal, and a third current path connected between said third terminal and a third current supply terminal, receives a third voltage at the other end of said third resistor as a reference potential, and outputs a fourth electric current independent of a temperature and corresponding to said fourth resistor from said third current supply terminal; and 
   a current mirror circuit which is connected to said high power supply voltage terminal to receive the high power supply voltage, and generates a bias current in accordance with an electric current supplied from a reference current terminal, 
   wherein said first, second, and third current supply terminals are connected to said reference current terminal. 
   According to one aspect of the present invention, there is provided a laser diode driving circuit comprising, 
   a sixth resistor having one end connected to said high power supply voltage terminal; 
   a fourth NPN bipolar transistor having a collector connected to the other end of said sixth resistor, and a base which receives one differential input signal; 
   a seventh resistor having one end connected to said high power supply voltage terminal; 
   a fifth NPN bipolar transistor having a collector connected to the other end of said seventh resistor, and a base which receives the other differential input signal; 
   a sixth NPN bipolar transistor having a collector connected to emitters of said fourth and fifth NPN bipolar transistors, a base connected to a current input terminal, and an emitter which is grounded either directly or via an eighth resistor; 
   a differential output unit which performs differential amplification by receiving the differential output signals, and generates a driving current signal for driving a laser diode from the collector of at least one of said fourth and fifth NPN bipolar transistors; 
   said bias current generating circuit; and 
   a driving current controller which receives the bias current generated by said bias current generating circuit, amplifies the received bias current, and supplies the amplified bias current to the current input terminal of said differential output unit. 
   According to one aspect of the present invention, there is provided an optical communication transmitter comprising, 
   said laser diode driving circuit; and 
   a laser diode which receives the driving current signal generated by said laser diode driving circuit. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a circuit diagram showing the arrangement of a bias current generating circuit according to an embodiment of the present invention; 
       FIG. 2  is a circuit diagram showing the arrangements of a laser diode driving circuit including the bias current generating circuit according to the same embodiment and an optical communication transmitter; 
       FIG. 3  is a graph showing the dependence of a bias current upon absolute temperature in the bias current generating circuit shown in  FIG. 1 ; 
       FIG. 4  is a circuit diagram showing an arrangement in which MOS transistors shown in  FIG. 1  are replaced with bipolar transistors; 
       FIG. 5  is a circuit diagram showing the arrangement of a laser diode driving circuit capable of using the bias current generating circuit of the present invention; 
       FIG. 6  is a circuit diagram showing the arrangement of a conventional bias current generating circuit; and 
       FIG. 7  is a graph showing the dependence of a bias current upon absolute temperature in the bias current generating circuit shown in  FIG. 6 . 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENT 
   An embodiment of the present invention will be described below with reference to the accompanying drawings. 
     FIG. 1  shows the arrangement of a bias current generating circuit according to the embodiment of the present invention. 
   Also, as shown in  FIG. 2 , a laser diode driving circuit LDDC according to the embodiment of the present invention comprises a driving current controller  11  including a bias current generating circuit BGC 11  shown in  FIG. 1 , a driver stage DA 100 , a differential output unit  2 , and a current source CS. 
   In addition, as shown in  FIG. 2 , an optical communication transmitter according to the embodiment of the present invention comprises the laser diode driving circuit LDDC according to this embodiment, and a laser diode LD. This optical communication transmitter further comprises an RC filter RCF which includes a resistor R 11 , capacitor C 1 , and resistor R 12  to suppress waveform distortion, a resistor Rd, and a choke coil CC. 
   The bias current generating circuit of this embodiment shown in  FIG. 1  is obtained by adding a low-potential-side current source circuit LCS 3  and high-potential-side current source circuit HCS 1  to the arrangement of the conventional bias current generating circuit shown in  FIG. 6 . In this embodiment, the same reference numerals as in the conventional circuit denote the same elements, and a detailed description thereof will be omitted. 
   The low-potential-side current source circuit LCS 3  has an operational amplifier OP 2 , NMOS transistor N 2 , and resistor R 5 . The non-inverting input terminal of the operational amplifier OP 2  receives, as a reference potential, a potential V 2  output from the output terminal of an operational amplifier OP 1  included in a bandgap reference circuit BGRC. The NMOS transistor N 2  has a drain connected to one terminal of a resistor R 6 , a source connected to the inverting input terminal of the operational amplifier OP 2 , and a gate connected to the output terminal of the operational amplifier OP 2 . The resistor R 5  is connected between the source of the transistor N 2  and the ground terminal. 
   The low-potential-side current source circuit LCS 3  described above and the resistor R 6  having one end connected to the drain of the transistor N 2  and the other end connected to a power supply voltage Vcc terminal form a voltage shift circuit VSC. The voltage shift circuit VSC shifts the voltage V 2  to a desired level and inputs the shifted voltage as a reference voltage V 3  to the non-inverting input terminal of an operational amplifier OP 3 . 
   The low-potential-side current source circuit LCS 3  is given the temperature-independent potential V 2  as a reference potential. Letting Ix denote an electric current which flows through the resistor R 6 , transistor N 2 , and resistor R 5  and is independent of the temperature, the electric current Ix is represented by
 
 Ix=V   2 / R   5   (4)
 
and the potential V 3  is represented by
 
 V   3 = Vcc−R   6 · Ix=Vcc −( R   6 / R   5 )· V   2   (5)
 
   The high-potential-side current source circuit HCS 1  has a resistor R 8  connected between the power supply voltage Vcc terminal and an external terminal PAD 3 , the operational amplifier OP 3  having an inverting input terminal connected to the external terminal PAD 3  and a non-inverting input terminal connected to the drain of the transistor N 2 , and a PMOS transistor P 1  having a source connected to the external terminal PAD 3 , a gate connected to the output terminal of the operational amplifier OP 3 , and a drain connected to the drain of a transistor N 3  of a low-potential-side current source circuit LCS 1 . 
   Since the temperature-independent reference voltage V 3  is supplied to the high-potential-side current source circuit HCS 1 , a temperature-independent electric current  14  flows through the transistor P 1 . The value of the electric current  14  is adjusted by the resistance value of the resistor R 8 . 
   This embodiment having the above arrangement operates as follows in accordance with the presence/absence of the external resistors R 7 , R 8 , and R 9 .
     (1) When the resistance values of the resistors R 7  and R 8  are infinite and that of the resistor R 9  is finite, the bias current Ibias maintains a constant value regardless of the temperature.   (2) When the resistance value of the resistor R 8  is infinite and those of the resistors R 7  and R 9  are finite, the bias current Ibias has a finite value at absolute zero and linearly increases with respect to the temperature.   (3) When the resistance values of the resistors R 8  and R 9  are infinite and that of the resistor R 7  is finite, the bias current Ibias is zero at absolute zero and proportional to the temperature.   (4) When the resistance value of the resistor R 9  is infinite and those of the resistors R 7  and R 8  are finite, the bias current Ibias maintains zero up to a certain finite temperature Tth and linearly increases above the temperature Tth.   

   The characteristics (1) to (3) are similar to those of the circuit shown in  FIG. 6 , but this embodiment additionally has the characteristic (4) described above. A graph of  FIG. 3  shows the characteristic (4). 
   As shown in  FIG. 3 , the electric current  13  flowing through a transistor P 2  as a mirror source of the bias current Ibias is zero until Tth (about 120K) and linearly increases above the temperature Tth. 
   At temperatures lower than the certain temperature Tth, the high-potential-side current source circuit HCS 1  for generating the electric current  14  does not function as a constant-current source and can supply only the same value as the electric current  11 . Therefore, the electric current  13  cannot be a negative electric current. 
   To raise the temperature Tth as a threshold value, the value of the resistance ratio R 8 /R 7  need only be decreased. 
   A rate R of increase of the bias current Ibias per degree of the temperature at a certain temperature T 0  is represented by
 
 R =1/( T   0 − T th)  (6)
 
   Accordingly, the change in bias current Ibias with temperature can be essentially unlimitedly increased by approaching the temperature Tth to T 0 . 
   In this embodiment as described above, the temperature Tth can be freely set by the values of the externally connected resistors R 7  and R 8 . Therefore, temperature compensation can be well performed even for a laser diode whose emission efficiency largely depends upon the temperature. 
   That is, the optical output power amplitude can be maintained constant regardless of the temperature even for a laser diode whose emission efficiency largely depends upon the temperature. 
   The above embodiment is merely an example and hence does not limit the present invention. Therefore, the embodiment can be variously modified within the spirit and scope of the present invention. 
   For example, in the above embodiment, transistors except for the two NPN bipolar transistors Q 1  and Q 2  included in the bandgap reference circuit BGRC are MOSFETs. However, as shown in  FIG. 4 , it is also possible to use NPN bipolar transistors instead of NMOS transistors, and PNP bipolar transistors instead of PMOS transistors. 
   Also, the current mirror circuit made up of the PMOS transistors P 2  and P 3  can be replaced with another circuit which performs a current mirror operation with higher accuracy. Furthermore, the bandgap reference circuit BGRC can have another arrangement instead of the circuit configurations shown in  FIGS. 1 ,  2 , and  4 . 
   As has been described above, in the bias current generating circuit according to the embodiment of the present invention, a first current supply terminal for supplying a first electric current dependent on the temperature and corresponding to a first resistor, a second current supply terminal for supplying a second electric current independent of the temperature and corresponding to a second resistor, and a third current supply terminal for supplying a third electric current independent of the temperature and corresponding to a third resistor are connected to the reference current terminal of the current mirror circuit, and a bias current is generated in accordance with an electric current supplied to this reference current terminal. 
   Also, the laser diode driving circuit and optical communication transmitter according to the embodiment of the present invention can well perform temperature compensation by supplying the bias current as described above to a laser diode, even when the temperature dependence of the emission efficiency of the laser diode is large, and can hold the optical output power amplitude constant regardless of the temperature.