Patent Publication Number: US-8125159-B2

Title: LED driving device with variable light intensity

Description:
PRIORITY CLAIM 
     The present application is a continuation of U.S. patent application Ser. No. 11/153,848, filed Jun. 14, 2005, which application claims the benefit of European Patent Application No. 04425437.3, filed Jun. 14, 2004; all of the foregoing application are incorporated herein by reference in their entireties. 
    
    
     TECHNICAL FIELD 
     Embodiments of the present disclosure relate to a LED driving device with variable light intensity. 
     BACKGROUND 
     As is known, thanks to the marked development of silicon-based technologies, high-efficiency light-emitting diodes (LEDs) are increasingly used in the field of lighting, whether industrial or domestic lighting. For example, high-efficiency LEDs are commonly used in automotive applications (in particular for the manufacturing the rear lights of motor vehicles), in road signs, or in traffic lights. 
     According to the light intensity that it is desired to obtain, it is possible to coupled alternately a number of LEDs in series or a number of arrays of LEDs in parallel (by the term array is meant, in this context, a certain number of LEDs coupled in series to one another). Clearly, the number of LEDs and the criterion of connection adopted determine the characteristics of the driving device (hereinafter “driver”) that must be used for driving the LEDs. 
     In particular, with the increase in the number of LEDs coupled in series, the value of the output voltage of the driver must increase, while, with the increase in the number of arrays in parallel, the value of the current that the driver must be able to furnish for supplying the LEDs must increase. 
     Furthermore, the intensity of current supplied to a LED determines its spectrum of emission and hence the color of the light emitted. It follows that, to prevent the spectrum of emission of a LED from varying, it is of fundamental importance that the supply current should be kept constant, and hence generally the driver used for driving the LEDs is constituted by a current-controlled DC/DC converter. 
     As is known, the topology of the DC/DC converter differs according to the type of application envisaged. Normally, the configurations “flyback” or “buck” are used, respectively, if an electrical insulation is required or if the driver is supplied directly by the electric power-supply mains (and hence there is no need to step up the input voltage), whereas the “boost” configuration is used when the driver is battery-supplied and it is hence necessary to step up the input voltage. 
     In many applications, it is required to vary the intensity of the light emitted by the LED gradually, this operation being known by the term “dimming”. 
     On the other hand, it is not possible to simply vary (either decrease or increase) the supply current supplied to the LED, in so far as it is not possible to accept the change of color of the emitted light (typically, constancy in the spectrum of emission is required), color which, as mentioned, depends upon the supply current. 
     For this reason, currently drivers for LEDs comprise a pulse-width-modulation (PWM) control for turning on and turning off LEDs at low-frequency (100-200 Hz), with a ratio between turning-on time and turning-off time (duty cycle) that is a function of the level of light intensity required. 
     To achieve turning-on and turning-off of LEDs, a switch is set in series between the output of the DC/DC converter and the LEDs themselves. Said switch, controlled in PWM, enables or disables the supply of the LEDs. In particular, during the ON phase of the PWM control signal, the switch closes, enabling passage of the supply current to the LEDs and hence their turning-on, while during the OFF phase of the PWM control signal the switch is open, interrupting passage of the supply current and hence causing turning-off of the LEDs. Clearly, the frequency of the PWM control signal is such that the human eye, given the stay time of the image on the retina, does not perceive turning-on and turning-off of the LEDs, since it perceives a light emitted in a constant way. 
     The circuit described, albeit enabling dimming of the LEDs to be obtained, presents, however, certain disadvantages linked to the presence of a switch coupled to the output of the DC/DC converter in series with the load. 
     In fact, in the majority of applications, high-efficiency LEDs require high supply currents, in the region of various hundreds of mA (typically between 100 mA and 700 mA). Consequently, the switch set in series to the load must be a power switch; moreover, it must have low leakages in conduction in order not to limit the efficiency for driving. On the other hand, the higher the supply current required by the LEDs, the more critical the choice of the power switch, and consequently the higher the cost of the switch and as a whole the cost of construction of the driver. 
     Embodiments of the present disclosure provide a LED-driving device that is free from the drawbacks described above, and in particular that enables adjustment of the light intensity of the LEDs in a more economical and efficient way. 
     SUMMARY 
     According to an embodiment of the present disclosure there is provided a LED driving device and method with variable light intensity. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a better understanding of the present disclosure, there is now described a preferred embodiment thereof, which is provided purely by way of non-limiting example and with reference to the attached drawings, wherein: 
         FIG. 1  is a block diagram of a LED driving circuit according to an embodiment of the present disclosure; 
         FIG. 2  shows time diagrams of some circuit quantities of the circuit of  FIG. 1 ; 
         FIG. 3  is a detailed circuit diagram of the driving circuit of  FIGS. 1 ; and 
         FIG. 4  is a circuit diagram of an enabling stage of the circuit of  FIG. 1 , according to a further embodiment of the present disclosure. 
     
    
    
     DETAILED DESCRIPTION 
     The following discussion is presented to enable a person skilled in the art to make and use the disclosure. Various modifications to the embodiments will be readily apparent to those skilled in the art, and the generic principles herein may be applied to other embodiments and applications without departing from the spirit and scope of the present disclosure. Thus, the present disclosure is not intended to be limited to the embodiments shown, but is to be accorded the widest scope consistent with the principles and features disclosed herein. 
     The idea underlying embodiments of the present disclosure draws its origin from the consideration that a LED can be considered as a normal diode, with the sole difference that it has a higher threshold voltage V f  (normally around 3 V as against the 0.7 V of a normal diode). It follows that a LED automatically turns off when it is biased with a voltage lower than the threshold voltage V f . In particular, to obtain turning-off of the LEDs, the driving circuit passes from a current control mode to a voltage control mode, which limits the output voltage to a value lower than the threshold voltage of the LEDs. By varying the intervals of time when the two control modes are active, for example via a PWM control, it is possible to vary the light intensity of the LEDs. 
     For a better understanding of the above, reference is now made to  FIG. 1 , which illustrates a LED-driving device  1 . 
     In detail, the driving device  1  comprises a pair of input terminals  2 ,  3 , receiving a supply voltage V in  (in this case, coming from the electric power-supply mains) and a first and a second output terminals  4 ,  5 , coupled to the load that must be driven. In particular the load is formed by 1 to N arrays  6  of LEDs  7  arranged in parallel, and each array  6  can contain a variable number of LEDs  7  coupled in series to each other. 
     The driving device  1  moreover comprises an AC/DC converter  8  coupled to the input terminals  2 ,  3  and operating as a rectifier of the mains voltage, and a supply stage  9 , cascade-coupled to the AC/DC converter  8  and supplying an output supply voltage V out  and an output supply current I out . The supply stage  9  is basically formed by a DC/DC converter and has a first and a second outputs  10   a ,  10   b , coupled to the first and the second output terminals  4 ,  5 , respectively. A current sensor  11  is coupled between the second output terminal  5  of the driving device  1  and the second output  10   b  of the supply stage  9 , and outputs a current-feedback signal V 1   fb  proportional to the current flowing in the load and co-operating with the supply stage  9  for controlling of the current I out . Typically, the current sensor  11  comprises a sensing resistor (as described in detail in  FIG. 3 ). 
     The driving device  1  moreover comprises a PWM control circuit  13 , of a known type, and an enabling stage  14 . The PWM control circuit  13  receives an external command, indicated schematically by the arrow  17 , and generates a PWM control signal, the pulse width whereof is modifiable via the external control circuit  13 , in a known way. 
     The enabling stage  14 , controlled by the PWM control signal, is coupled between the first and second outputs  10   a ,  10   b  of the supply stage  9  and outputs a voltage-feedback signal V 2   fb  having two functions: on the one hand, it enables/disables the voltage control of the supply stage  9 ; on the other, it supplies an information correlated to the voltage V out . 
     To this end, the enabling stage  14  comprises a voltage sensor formed by a resistive divider (as illustrated in detail in  FIG. 3 ), the output signal whereof forms the voltage-feedback signal V 2   fb . In this way, in the voltage-control mode, the supply stage  9  can limit the output voltage V out  to a value smaller than the threshold voltage of the arrays  6 , equal to the sum of the threshold voltages of the LEDs  7  in each array  6 . If the arrays  6  contain a different number of LEDs  7 , the output voltage V out  is limited to a value smaller than the minimum total threshold value of the arrays  6 . For example, if even just one array  6  is made up of a single LED  7 , the output voltage V out  is limited to a value smaller than the threshold voltage V f  of a LED; for example it can be set at the non-zero value of 2 V. 
     Operation of the driving device  1  is as follows. 
     In normal operating conditions, when the voltage control of the supply stage  9  is disabled by the enabling stage  14  (for example, during the OFF phase of the PWM control signal), the supply stage  9  works in a current control mode and uses the current-feedback signal V 1   fb  so that the output current I out  has a preset value, such as to forward bias the LEDs  7 , which thus conduct and emit light. 
     In particular, the output current I out  has a value equal to the sum of the currents I 1 , . . . I N  that are to be supplied to the various arrays  6  for forward biasing the LEDs  7 . The output voltage V out  has, instead, a value fixed automatically by the number of driven LEDs  7  (for example, a total threshold voltage value of 35 V, when an array  6  is made up of ten LEDs and each LED has an on-voltage drop of 3.5 V). 
     In this step, then, the current control enables precise control of the value of the supply current of the LEDs  7  according to the desired spectrum of emission. 
     When, instead, the voltage control of the supply stage  9  is enabled by the enabling stage  14  (in the example, during the ON phase of the PWM control signal), the value of the voltage V out  is limited to a value smaller than the minimum threshold voltage of the arrays  6 , so causing turning-off of the LEDs  7 , as explained in greater detail with reference to  FIG. 3 . 
     The PWM control circuit  13 , by varying appropriately the duty cycle of the PWM control signal that controls the enabling stage  14 , enables regulation of the intensity of the light emitted by the LEDs  7 . In the example, with the increase in the duty cycle, the time interval when the control of the supply stage  9  is a current control and the LEDs  7  are forward biased, increases, and consequently the intensity of the light emitted increases. In particular, a duty cycle equal to zero corresponds to a zero light intensity, while a duty cycle equal to one corresponds to a maximum intensity of the light emitted by the LEDs  7 . 
       FIG. 2  shows the time plots of the PWM control signal generated by the PWM control circuit  13 , of the output current I out , and of the output voltage V out  during normal operation of the driving device  1 . 
     As may be noted, during the ON phase of the PWM control signal the supply stage  9  works in a current control mode, outputting the current I out  for supply of the LEDs  7 ; the voltage V out  assumes a value, for example 35 V. Instead, during the OFF phase of the PWM control signal the supply stage  9  works in a voltage control mode, limiting the output voltage V out  to a value, for example 2 V, while the current I out  goes to zero. 
     By appropriately varying the duty cycle of the PWM control signal (as indicated by the arrows in  FIG. 2 ), it is possible to regulate appropriately the level of light intensity of the LEDs  7 . 
       FIG. 3  shows a possible circuit embodiment of the driving device  1 , when the driving device  1  is supplied by the electrical power mains and a galvanic insulation is moreover required. 
     In particular, a detailed description of the current sensor  11 , the enabling stage  14 , and the supply stage  9  is given, since the other components are of a known type. 
     In detail, the current sensor  11  comprises a sensing resistor  20  coupled between the second output  10   b , which is grounded, of the supply stage  9  and the second output terminal  5 . 
     The enabling stage  14  comprises a first resistor  27  and a second resistor  28 , coupled in series. The first resistor  27  is coupled between the first output terminal  4  and a first intermediate node  31 , while the second resistor  28  is coupled between the first intermediate node  31  and a second intermediate node  32 . The voltage-feedback signal V 2   fb  is present on the first intermediate node  31 . The enabling stage  14  further comprises a third resistor  37  coupled between the second intermediate node  32  and the second output  10   b  of the supply stage  9 , and a bipolar transistor  40  of an NPN type, having its collector terminal coupled to the second intermediate node  32 , its emitter terminal coupled to the second output  10   b , and its base terminal receiving the PWM control signal generated in a known way by the PWM control circuit  13 . The third resistor  37  forms, together with the first resistor  27  and the second resistor  28 , a resistive divider  12 , controllable via the PWM control signal. 
     The supply stage  9  comprises a DC/DC converter  15 , of a “flyback” type, cascaded to the AC/DC converter  8  and having the first output  10   a  and the second output  10   b . The supply stage  9  moreover comprises a selection stage  16  receiving the current-feedback signal V 1   fb  and the voltage-feedback signal V 2   fb , and having an output coupled to a feedback input  26  of the DC/DC converter  15 . In particular, the selection stage  16  alternately feeds the feedback input  26  with the voltage-feedback signal V 2   fb  and the current-feedback signal V 1   fb  so as to enable, respectively, voltage control and current control. 
     In detail, the selection stage  16  comprises a first and a second operational amplifiers  21 ,  30 . The first operational amplifier  21  has its inverting terminal coupled to the second output terminal  5  and receiving the current-feedback signal V 1   fb , its non-inverting terminal receiving a first reference voltage V ref1 , of preset value, and an output coupled, via the interposition of a first diode  24 , to a feedback node  23 , which is in turn coupled to the feedback input  26  of the DC/DC converter  15 . The first diode  24  has its anode coupled to the output of the first operational amplifier  21  and its cathode coupled to the feedback node  23 . Furthermore, a first capacitor  25  is coupled between the inverting terminal of the first operational amplifier  21  and the cathode of the first diode  24 . The second operational amplifier  30  has its inverting terminal coupled to the first intermediate node  31  and receiving the voltage-feedback signal V 2   fb , its non-inverting terminal receiving a second reference voltage V ref2 , of preset value, and an output coupled to the feedback node  23  via a second diode  34 . The second diode  34  has its anode coupled to the output of the second operational amplifier  30  and its cathode coupled to the feedback node  23 . Furthermore, a second capacitor  35  is coupled between the inverting terminal of the second operational amplifier  30  and the cathode of the second diode  34 . 
     In practice, two distinct feedback paths are formed, which join in the feedback node  23 . A first path, which comprises the current sensor  11 , enables current control through the current-feedback signal V 1   fb , in so far as it detects the value of the output current I out  via the sensing resistor  20 . A second path, which comprises the enabling stage  14 , enables, instead, voltage control through the voltage-feedback signal V 2   fb , in so far as it detects the value of the output voltage V out  via the resistive divider  12 . 
     The two feedback paths are enabled alternately by the enabling stage  14 . 
     In fact, the transistor  40  acts as a switch controlled by the PWM control signal generated by the PWM control circuit  13 , determining, with its opening and its closing, two different division ratios of the resistive divider  12  and hence different values of the voltage-feedback signal V 2   fb . 
     In detail, when the transistor  40  is turned on (ON phase of the PWM control signal), the third resistor  37  is short-circuited and the resistive divider  12  is formed only by the first resistor  27  and second resistor  28  having resistances R 1  and R 2 , respectively. In this situation, the voltage-feedback signal V 2   fb  assumes a first value V 2   fb1  equal to 
               V   ⁢           ⁢     2     fb   ⁢           ⁢   1         =       V   out     ·       R   2         R   2     +     R   1                 
whereas, when the transistor  40  is turned off (OFF phase of the PWM control signal), the resistive divider  12  is formed by the first resistor  27 , the second resistor  28 , and a third resistor  37 , wherein the third resistor  37  has a resistance R 3 . In this case, the voltage-feedback signal V 2   fb  assumes a second value V 2   fb2  equal to
 
               V   ⁢           ⁢     2     fb   ⁢           ⁢   2         =       V   out     ·         R   2     +     R   3           R   2     +     R   3     +     R   1                 
where obviously V 2   fb2 &gt;V 2   fb1 .
 
     It follows that, during the ON phase of the PWM control signal, the inverting terminal of the second operational amplifier  30  is at a potential V 2   fb1  smaller than that of the non-inverting terminal receiving the second reference voltage V ref2 , so that the output of the second operational amplifier  30  becomes positive, causing an off-state of the second diode  34 . Instead, the first operational amplifier  21  receives, on its inverting terminal, a voltage V 1   fb  proportional to the current flowing in the sensing resistor  20 , greater than the first reference voltage V ref1 , and hence the first diode  24  is on. In this way, the feedback node  23  is coupled to the first feedback path, and the voltage control is disabled, whereas the current control through the current sensor  11  is enabled. The first reference voltage V ref1  has a low value (for example, 100 mV) so as to limit the power dissipation on the sensing resistor  20 . 
     Instead, during the OFF phase of the PWM control signal, the inverting terminal of the second operational amplifier  30  is at a potential V 2   fb2  higher than that of the non-inverting terminal, receiving the second reference voltage V ref2 , so that the output of the second operational amplifier  30  becomes negative, causing turning-on of the second diode  34 . Instead, in this situation, the first diode  24  is turned off. In this way, the feedback node  23  is coupled to the second feedback path, and consequently the voltage control is enabled, which limits the output voltage V out  to a value lower than the threshold voltage of the array  6 , as described above. The value of the second reference voltage V ref2  supplied to the non-inverting terminal of the second operational amplifier  30 , and the values of the resistances are chosen so that the output voltage V out  assumes the desired value. 
     The driving device described herein presents the following advantages, although all such as advantages need not be realized by all embodiments of the present disclosure. 
     First, it has a driving efficiency greater than known driving devices, in so far as it does not have elements arranged in series to the load that generate leakages. 
     Furthermore, the production costs are decidedly lower, in so far as the need for the presence of a costly power switch is avoided, since the latter is replaced by a simple signal switch, of negligible cost. 
     Finally, in the case of integration of the driving device, it does not present problems of power dissipation, with consequent savings and greater simplicity of production. 
     Finally, it is clear that modifications and variations can be made to the device for driving LEDs described and illustrated herein, without thereby departing from the scope of the present disclosure, as defined in the annexed claims. 
     In particular, it is emphasized that the present driving device, although designed for driving arrays of LEDs of the type described, does not include said light-emitting elements, which consequently do not form part of the driving device. 
     Furthermore,  FIG. 4  shows a further embodiment of the enabling stage  14  of the driving device  1 . In particular, the resistive divider of the enabling stage  14  comprises only the first resistor  27  and the second resistor  28 , the first resistor  27  being coupled between the first output  10   a  and the first intermediate node  31 , and the second resistor  28  being coupled between the first intermediate node  31  and the second intermediate node  32 . The bipolar transistor  40  still has its collector terminal coupled to the second intermediate node  32 , its emitter terminal coupled to the second output  10   b , and its base terminal receiving the PWM control signal generated by the PWM control circuit  13 . According to this further embodiment, the enabling stage  14  further comprises a zener diode  42 , which is coupled between the first intermediate node  31  and ground of the driving device  1 . 
     Operation of the driving device  1  according to this further embodiment is now described, referring to the situation in which the driving device  1  drives an array  6  having a number of LEDs  7  equal to N led . 
     When the transistor  40  is turned on (ON phase of the PWM control signal), the voltage-feedback signal V 2   fb  assumes the first value V 2   fb1 : 
     
       
         
           
             
               V 
               ⁢ 
               
                   
               
               ⁢ 
               
                 2 
                 
                   fb 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   1 
                 
               
             
             = 
             
               
                 V 
                 out 
               
               · 
               
                 
                   R 
                   2 
                 
                 
                   
                     R 
                     2 
                   
                   + 
                   
                     R 
                     1 
                   
                 
               
             
           
         
       
     
     The first value V 2   fb1  is smaller than the second reference voltage V ref2 , so that the current control through the current sensor  11  is enabled (as previously described). The LEDs  7  are thus in the on-state and the output voltage V out  is N led ⊚3.5 V (3.5 V being the on-voltage drop of each LED  7  of the array  6 ). 
     Instead, during the OFF phase of the PWM control signal, the transistor  40  is turned off, and the voltage-feedback signal V 2   fb  is instantaneously pulled up to a value higher than the second reference voltage V ref2  (zener diode  42  can limit this value so that a maximum voltage that can be applied to the second operational amplifier  30  is not exceeded), thus enabling voltage control. Therefore, the output current I out  flowing in the LEDs  7  falls to zero, while the output voltage V out  decreases down to N led ⊚2 V (2 V being the threshold voltage of each LED  7 ). Further decrease of the output voltage V out  is not possible, due to high output impedance. 
     Capacitor C at the output of the supply stage  9  thus experiences a voltage variation ΔV at the switching between the ON and the OFF phase of the PWM control signal, which is equal to N led ⊚1.5V. This voltage variation ΔV causes a delay t in the reactivation of LEDs  7  (due to the charging of capacitor C) of: 
     
       
         
           
             t 
             = 
             
               
                 
                   
                     C 
                     
                       I 
                       out 
                     
                   
                   · 
                   Δ 
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 V 
               
               = 
               
                 
                   C 
                   
                     I 
                     out 
                   
                 
                 · 
                 
                   ( 
                   
                     1.5 
                     · 
                     
                       N 
                       led 
                     
                   
                   ) 
                 
               
             
           
         
       
     
     Given a same value of the capacitor C, the delay t in this further embodiment is greatly reduced with respect to the circuit shown in  FIG. 3 . In fact, in the circuit of  FIG. 3  the voltage variation ΔV is:
 
Δ V =(3.5 ·N   led −2)
 
since the output voltage V out  is limited to 2 V during the OFF stage of the PWM control signal (irrespective of the number of LEDs  7 ), and so the delay t is given by:
 
             t   =           C     I   out       ·   Δ     ⁢           ⁢   V     =       C     I   out       ·     (       3.5   ·     N   led       -   2     )               
In particular, the advantage in terms of reduction of the delay time t increases with the increase of the number N led  of LEDs  7  in the array  6 .