Patent Publication Number: US-7903044-B2

Title: Dielectrically-loaded antenna

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims the benefit of the filing date of U.S. Provisional Patent Application Ser. No. 60/920,928 filed on Mar. 30, 2007, the entire disclosure of which is incorporated herein by reference. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to a dielectrically-loaded antenna for operation at frequencies in excess of 200 MHz and a portable wireless terminal incorporating such an antenna. 
     BACKGROUND OF THE INVENTION 
     Such antennas are disclosed in a number of patent publications of the present applicant, including GB2292638A, GB2309592A, GB2310543A, GB2338605A, GB2346014A GB2351850A and GB2367429A. Each of these antennas has at least one pair of diametrically opposed helical antenna elements which are plated on a substantially cylindrical electrically insulative core made of a material having a relative dielectric constant greater than 5. The material of the core occupies the major part of the volume defined by the core outer surface. Extending through the core from one end face to an opposite end face is an axial bore containing a coaxial feed structure that comprises an inner conductor surrounded by a shield conductor. At one end of the bore the feed structure conductors are connected to respective antenna elements which have associated connection portions adjacent the end of the bore. At the other end of the bore, the shield conductor is connected to a conductor which links the antenna elements and, in each of these examples, is in the form of a conductive sleeve encircling part of the core to form a balun. Each of the antenna elements terminates on a rim of the sleeve and each follows a respective helical path from its connection to the feed structure. 
     Some of the above prior patent publications disclose quadrifilar helical antennas intended primarily for receiving or transmitting circularly polarised electromagnetic waves. Each of these antennas has four helical tracks plated on the cylindrical surface of the core, or four groups of helical tracks, each group forming a composite antenna element and comprising two tracks separated by a narrow slit. 
     Whether the antenna has four helical antenna elements or two, the connection portions connecting the antenna elements to the feed structure conductors are radial tracks plated on a planar end surface of the core. 
     It is known to provide a quadrifilar helical antenna with an impedance matching network. This may be embodied as a small printed circuit or laminate board secured to the top end face of the core where it provides coupling between the feed structure and radial connection portions such as those disclosed in the above mentioned prior patent publications. An antenna having such a matching network is disclosed in our co-pending U.S. patent application Ser. No. 11/472,587. The disclosures of this application and each of the prior patent publications referred to above are specifically incorporated in this specification by reference. 
     It is an object of the invention to provide an improved dielectrically-loaded antenna. 
     SUMMARY OF THE DISCLOSURE 
     According to a first aspect of the present disclosure, a dielectrically-loaded antenna having an operating frequency in excess of 200 MHz comprises: an electrically insulative core of a solid material that has a relative dielectric constant greater than 5 and occupies the major part of the interior volume defined by the core outer surface, and a three-dimensional antenna element structure that is on or adjacent the core outer surface and comprises at least six elongate conductive elongate conductive antenna elements. The antenna elements are typically substantially axially coextensive and substantially uniformly spaced apart around an axis of the antenna, the elements being arranged in pairs, each juxtaposed so as to be on diametrically opposite sides of the axis. The antenna is resonant in a circularly polarised mode of resonance at the operating frequency, the resonant mode being characterised by a rotating dipole, and voltage maxima being excited on each of the elongate antenna elements in succession in the direction of rotation. 
     The preferred antenna includes a pair of antenna element coupling nodes, each of the said pairs of elements having one antenna element connected to one of the coupling nodes and another antenna element connected to the other coupling node. The preferred antenna also has a common interconnecting conductor for the elongate antenna elements, advantageously in the form of a conductive ring interconnecting ends of the elongate conductive elements. This conductor may encircle the axis and lie generally in plane extending perpendicularly to the axis. Preferably, this interconnecting conductor encircles the core on the outer side surface portion of the latter and defining a conductive path around the core. Each elongate antenna element has a first end connected to one or other of the coupling nodes and a second end connected to the common interconnecting conductor, the connections of the second ends being at equally spaced connection points. 
     Advantageously the electrical length of the conductive path formed by the common interconnecting conductor is substantially equal to a whole number (1, 2, 3, . . . ) of guide wavelengths corresponding to the operating frequency of the antenna. This enhances the circularly polarised resonant mode of the antenna since the common interconnecting conductor has a ring resonance at the operating frequency, promoting the progression of the rotating dipole around the uniformly spaced-apart elongate antenna elements. 
     The common interconnecting conductor may be a narrow annular conductive track, both edges of which are on the outer side surface portion of the core. Such a configuration is particularly suitable for an endfire multifilar helical antenna. Alternatively, the common interconnecting conductor may be constituted by a conductive sleeve that surrounds the core and extends over an end surface portion to make a connection with the shield conductor of a coaxial transmission line feeder. This feeder passes through the core to connections with the elongate antenna elements at an opposite end surface portion of the core. Such a sleeve may form an integral balun, as described in the above-referenced prior patent applications of the present applicant. 
     The ends of the elongate antenna elements are preferably equiangularly spaced around the central axis, the physical spacings being equal to the differences in phase between voltages and currents on the respective elements. In general, the physical angular spacing between successive elongate antenna elements does not vary by more than 2:1, at both the ends of the helices and at locations between their ends. 
     In one embodiment of the invention, the elongate antenna elements are helices of substantially equal length. Especially with a common interconnecting conductor exhibiting a ring resonance at the operating frequency, phasing of the currents and voltages in the elongate antenna elements is not entirely dependent on the electrical lengths of such elements. In other embodiments, however, phasing of the elements may be achieved by arranging for the common interconnecting conductor to follow a non-planar path, so that the above-mentioned second ends of the antenna elements of each group of elements, which have their first ends connected to a respective one of the coupling nodes, are at different distances from the first ends, the nature of the variation of such distances in a given direction of rotation about a central axis of the antenna depending on the arrangement of connections between the first ends of the elements and the respective coupling nodes. It has been found, in particular, that edge effects in the arrangement of the connections tend to favour a non-monotonic progression of element lengths so that, for instance, elements which constitute inner elements of their respective groups are longer than the outer elements. However, it is also possible for the second ends of the antenna elements of each group to be progressively closer to the first ends in a given direction of rotation about a central axis of the antenna. In particular, the conductive path provided by the common interconnecting conductor may be inclined or progressively stepped in a first direction between each of the antenna elements of each group of such elements connected at their first ends to a respective one of the coupling nodes, and inclined or stepped in the opposite direction between the groups. Thus, in this instance, the conducting path may be regarded as having two peaks and two troughs, the peaks and troughs occurring alternately and the slopes between the peaks and troughs being such that the two slopes in the first direction are much less steep than the two slopes in the opposite direction. 
     In one preferred embodiment each elongate antenna element has its first end coupled to its respective coupling node and its second end spaced from the first end, the element being dimensioned to yield a predetermined electrical path length between the respective coupling node and the second end. The elongate antenna elements coupled to each node form a group of neighbouring elements which are arranged so as to be angularly spaced apart with respect to the axis and such that their respective electrical path lengths differ and thereby form a monotonic progression, the sense of the progression being the same for each group. 
     The second ends of the antenna elements are preferably linked. Thus, in the preferred embodiments, each elongate antenna element of each pair of such elements has a first end coupled to a respective one of the coupling nodes and the second end which is linked to the second end of the other elongate antenna element of the pair to form at least part of a conductive loop that is generally symmetrical about the axis and that has a predetermined resonant frequency. The loops formed by such pairs of elongate antenna elements are angularly distributed with respect to the axis, the respective resonant frequencies of the loops varying monotonically with angular orientation about the axis. In such a case, the second ends of the elongate antenna elements may be linked by the common interconnecting conductor encircling the core, such that their second ends are defined by the connections of the elements to a common annular edge of the interconnecting conductor, which edge, in terms of its axial position, varies in height non-monotonically across each group of elongate antenna elements. 
     It will be noted that in preferred embodiments of the invention phasing of currents and voltages on the helical antenna elements is achieved by conductors on the core, rather than using an external network. 
     The preferred embodiments of the invention each take the form of an octafilar helical antenna having four pairs of elongate helical antenna elements on a cylindrical surface portion of the core, the angular spacing of neighbouring such elements being 45° at the cylinder axis. Preferably, each helical element executes a half turn about the axis, although quarter-turn elements may be used. In general, the elements may execute M/4 turns where M is an integer (1, 2, 3 . . . ). 
     The helical elements preferably comprise conductive tracks on the core outer side surface portion. They may be pure helices or they may deviate from a pure helical path, e.g. by being meandered. It is also possible to alter their electrical length, in each case, by meandering, for instance, only one of the edges, or by meandering the two edges of the track to different amplitudes. It is to be noted that efficiency of the antenna is greater than that associates with an equivalent quadrifilar antenna because the number of conductive track edges of the radiating structure is greater. At typical operating frequencies of such antennas, currents tend to be confined to the edges or peripheries of conductors. It follows that increasing the number of edges connected in parallel reduces ohmic losses and hence increases efficiency. This gain in efficiency can be used to yield improved sensitivity for receiving equipment and greater effective transmitted power for transmitting equipment. Alternatively, it can be used to provide antennas which are smaller than prior antennas of a given efficiency. Thus, for instance, where dielectrically loaded antennas of 10 mm diameter have been used for a predetermined purpose, antennas such as those described in this specification can be made with a diameter of 7.5 mm, with no significant loss in efficiency. In such a case the relative dielectric constant of the core material is typically in the range of from 50 to 100. 
     In one embodiment of the invention, the elongate antenna elements connected to each of the above-mentioned coupling nodes comprise a group of antenna elements spaced apart laterally with respect to each other and having two outer elements and at least one inner element between the outer elements of the group, the or each inner element having a greater electrical length than the outer element. Such a configuration is particularly applicable to an antenna in which radial connection portions on an end surface of the core are used to connect the coupling nodes to the respective elongate antenna elements. 
     In the case where the antenna element structure comprises an odd number of pairs of elongate antenna elements, each group of elements, coupled to a respective coupling node, has a middle antenna element and outer elements. The middle antenna element has an associated resonance at a frequency which is midway between the frequencies of resonances associated respectively with the antenna elements of the group which are on each respective side of the middle element. It has been found that an odd number of pairs of elongate conductive antenna elements has advantages in terms of bandwidth, especially in the case of an antenna having six helical antenna elements. 
     In preferred embodiments of the invention, the relative dielectric constant of the core material is greater than 10 and, more preferably, greater than 20. 
     According to a second aspect of the present disclosure, the portable wireless communication terminal includes an antenna as described above and a generally planar circuit board having an electrically conductive layer, wherein such a layer has an edge adjacent the antenna element structure of the antenna and extends generally radially outwardly from the core of the antenna with respect to the antenna axis. The conductive layer may lie in a plane generally parallel to the axis or in a plane containing the axis. Such an arrangement is advantageous in terms of the detuning effect of the electrically conductive layer of the circuit board on the antenna, the presence of at least three pairs of elongate conductive antenna elements being less susceptible to detuning than those of a multifilar helical antenna having fewer elements to the extent that the conductive layer can typically extend relatively close to the antenna surface over the axial length of the elements, e.g. typically significantly closer than 3 mm and, in the preferred communication terminal in accordance with the invention, to within 1 mm of the antenna element structure. 
     The invention will now be described below by way of example with reference to the drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       In the drawings: 
         FIG. 1  is a perspective view of a first antenna in accordance with the invention; 
         FIG. 2  is a perspective view of a plated antenna core of the antenna of  FIG. 1 , viewed from a distal end and one side; 
         FIG. 3  is an axial cross-section of a feed structure of the antenna of  FIG. 1 ; 
         FIG. 4  is a representation of the conductor pattern on the outer cylindrical surface of the antenna of  FIG. 1 , transformed to a plane; 
         FIG. 5  is a similar representation of an alternative conductor pattern; 
         FIG. 6  is a detail of the feed structure shown in  FIG. 4 , showing a laminate board thereof detached from a distal end portion of a feeder transmission line; 
         FIGS. 7A ,  7 B and  7 C are diagrams showing conductor patterns of three conductive layers of the laminate board of the feeder structure; 
         FIG. 8  is a perspective view of a second antenna in accordance with the invention; 
         FIG. 9  is a see-through representation of a conductor pattern of the antenna of  FIG. 8 ; 
         FIG. 10  is a representation of the conductor pattern on the outer cylindrical surface of the antenna of  FIG. 8 , transformed to a plane; 
         FIG. 11  is a perspective view of an assembly comprising a third antenna in accordance with the invention and a printed circuit board providing a balun and front-end receiver circuitry; 
         FIG. 12  is an axial cross-section of the printed circuit board of the assembly of  FIG. 11  and part of the antenna to which it is mounted; 
         FIG. 13  is a diagrammatic perspective view of a portable wireless terminal in accordance with the invention; 
         FIG. 14  is a perspective view of a second antenna in accordance with the invention; 
         FIG. 15  is a diagram illustrating a balun rim profile of the antenna of  FIG. 14 ; and 
         FIG. 16  is a graph illustrating individual frequency responses of conductor tracks of the antenna of  FIG. 14 . 
     
    
    
     DETAILED DESCRIPTION OF THE DRAWINGS 
     Referring to  FIGS. 1 and 2 , an octafilar helical antenna in accordance with the invention has an antenna element structure with eight elongate antenna elements in the form of eight axially coextensive helical conductive tracks  10 A,  10 B,  10 C,  10 D,  10 E,  10 F,  10 G,  10 H plated or otherwise metallised on the cylindrical outer surface of a cylindrical core  12 . The core is made of a ceramic material. In this case it is a barium titanate material having a relative dielectric constant of in the region of  36 . This material is noted for its dimensional and electrical stability with varying temperature. Dielectric loss is generally negligible. In this embodiment, the core has a diameter of 10 mm. The length of the core is greater than the diameter but, in other embodiments of the invention, it may be less. The core is produced by pressing, but may be produced in an extrusion process, the core then being fired. 
     This preferred antenna is a backfire helical antenna in that it has a coaxial transmission line housed in an axial bore  12 B that passes through the core from a distal end face  12 D to a proximal end face  12 P of the core. Both end faces  12 D,  12 P are planar and perpendicular to the central axis of the core. They are oppositely directed, in that one is directed distally and the other proximally in this embodiment of the invention. The coaxial transmission line is a rigid coaxial feeder which is housed centrally in the bore  12 B with the outer shield conductor spaced from the wall of the bore  12 B so that there is, effectively, a dielectric layer between the shield conductor and the material of the core  12 . Referring to  FIG. 3 , the coaxial transmission line feeder has a conductive tubular outer shield  16 , a first tubular air gap or insulating layer  17 , and an elongate inner conductor  18  which is insulated from the shield by the insulating layer  17 . The shield  16  has outwardly projecting and integrally formed spring tangs  16 T or spacers which space the shield from the walls of the bore  12 B. A second tubular air gap exists between the shield  16  and the wall of the bore  12 B. The insulative layer  17  may, instead, be formed as a plastics sleeve, as may the layer between the shield  16  and the walls of the bore  12 B. At the lower, proximal end of the feeder, the inner conductor  18  is centrally located within the shield  16  by an insulative bush  18 B, as described in our co-pending U.S. patent application Ser. No. 11/472,587. 
     The combination of the shield  16 , inner conductor  18  and insulative layer  17  constitutes a transmission line of predetermined characteristic impedance, here 50 ohms, passing through the antenna core  12  for coupling distal ends of the antenna elements  10 A to  10 H to radio frequency (RF) circuitry of equipment to which the antenna is to be connected. The couplings between the antenna elements  10 A to  10 H and the feeder are made via conductive connection portions associated with the helical tracks  10 A to  10 H, these connection portions being formed as radial tracks  10 AR,  10 BR,  10 CR,  10 DR,  10 ER,  10 FR,  10 GR,  10 HR plated on the distal end face  12 D of the core  12 . Each connection portion extends from a distal end of the respective helical track to one of two arcuate conductors  10 AD,  10 EH plated on the core distal face  12 D adjacent the end of the bore  12 B. 
     The two arcuate conductors  10 AD,  10 EH are connected, respectively, to the shield and inner conductors  16 ,  18  by conductors on a laminate board  19  secured to the core distal face  12 D, as will described hereinafter. The coaxial transmission line feeder and the laminate board  19  together comprise a unitary feed structure before assembly into the core  12 , and their interrelationship may be seen by comparing  FIGS. 1 ,  2  and  3 . 
     Referring to  FIG. 3 , the inner conductor  18  of the transmission line feeder has a proximal portion  18 P which projects as a pin from the proximal face  12 P of the core  12  for connection to the equipment circuitry. Similarly, integral lugs (not shown) on the proximal end of the shield  16  project beyond the core proximal face  12 P for making a connection with the equipment circuitry ground. 
     The proximal ends of the antenna elements  10 A- 10 H are connected to a common virtual ground conductor  20 . In this embodiment, the common conductor is annular and in the form of a plated sleeve surrounding a proximal end portion of the core  12 . This sleeve  20  is, in turn, connected to the shield conductor  16  of the feeder by a plated conductive covering (not shown) of the proximal end face  12 P of the core  12 . 
     The eight helical antenna elements  10 A- 10 H constitute four pairs  10 A,  10 E;  10 B,  10 F;  10 C;  10 G;  10 D,  10 H of such elements, each pair having one helical element coupled to one of the arcuate conductors  10 AD,  10 EH and another element coupled to the other of the arcuate conductors  10 EH,  10 AD, and thence, respectively, to the inner conductor  18  and shield  16  of the transmission line feeder. In effect, therefore, the eight helical antenna elements  10 A- 10 D may be regarded as being arranged in two groups of four  10 A- 10 D,  10 E- 10 H, all of the elements  10 A- 10 D of one group being coupled to the first arcuate conductor  10 AD and all of the elements  10 E- 10 H of the other group being coupled to the second arcuate conductor  10 EH. Thus, the two arcuate conductors constitute first and second coupling nodes that interconnect the respective helical antenna elements, and provide common connections for the elements of each group to one or other of the conductors of the transmission line feeder. 
     In this preferred embodiment of the invention, the eight helical antenna elements  10 A- 10 H are of different lengths. More specifically, each pair of laterally opposed elements  10 A,  10 E;  10 B,  10 F;  10 C,  10 G;  10 D,  10 H has two elements of the same length, those of a first pair  10 A,  10 E being of a first length, those of a second pair  10 B,  10 F being of a second length, those of a third pair  10 C,  10 G being of a third length and those of the remaining fourth pair  10 D,  10 H being of a fourth length. It has been found that, in order to obtain a progression of individual resonant frequencies, one of the outermost elements of each group, in this case elements  10 A and  10 E, should be shorter than the others. Preferably, the physical length of the other elements  10 B,  10 F;  10 C,  10 G;  10 D,  10 H are substantially equal rather than being progressively greater in order to compensate for different path lengths in the conductive connection portions on the distal end face  12 D of the core  12 , i.e. the radial tracks  10 AR- 10 HR and arcuate conductors  10 AD,  10 EH, best seen in  FIG. 2 , and associated edge effects at the frequency of operation of the antenna. These differences in length are achieved by arranging for the rim  20 U of the sleeve  20  to be non-planar inasmuch as its distance from the distal end face  20 D of the core varies according to the angular position around the central axis of the core. This is seen most clearly in the diagram of  FIG. 4  which is a representation of the conductor pattern on the cylindrical outer surface of the core  12  shown as if the cylindrical surface has been transformed to a flat surface. In this transformed representation, each helical element  10 A- 10 H appears as a straight conductor track. As will be seen, the sleeve  20  has a rim  20 U with four inclined portions and two flat portions. These portions comprise a first portion  20 UAB inclined in a first direction, a second flat portion  20 UBD which is generally parallel to the edge of the distal end  12 D ( FIG. 2 ) and joined to the first portion, a third portion  20 UDE inclined in a second direction and joined to the second portion, a fourth portion  20 UEF again inclined in the first direction and joined to the third portion, a fifth portion  20 UFH parallel to the distal end  12 D and, to complete the annulus around the core, a sixth portion  20 UHA inclined in the second direction and joining the fifth and first portions. By connecting the proximal ends of the elements  10 A- 10 D of the first group to the rim at equally spaced intervals, as shown in  FIG. 4 , their lengths are varied as described above. The identical juxtaposition of the proximal ends of the elements  10 E- 10 H of the second group and the rim yields the same length variations. It will be appreciated that, instead of following an inclined slope, each inclined rim portion may instead be stepped to achieve the same length differences in the helical elements  10 A- 10 H. 
     In an alternative octafilar antenna in accordance with the invention, the lengths of the antenna elements of each group may vary monotonically with rotation around the core  12 . Such an arrangement is shown in  FIG. 5 . In this case, the rim  20 U has four inclined portions comprising a first portion  20 UAD inclined in a first direction, a second portion  20 UDE inclined in a second direction and joined to the first portion, a third portion  20 UEH inclined in the first direction and joined to the second portion, and, to complete the annulus around the core, a fourth portion  20 UHA inclined in the second direction and joining the third and first portions. In this way, the length of the elements  10 A- 10 D of the first group are made progressively greater in sequence. Similarly, the identical juxtaposition of the proximal elements of the elements  10 E- 10 H of the second group and the third rim portion also yield a progressively increasing element length. It will be evident that, in this case, the slope of the second and fourth rim portions  20 UDE,  20 UHA is steeper than the slopes of the first and third portions  20 UAD,  20 UEH owing to the uniform spacing of the helical elements  10 A- 10 H around the core  12 . 
     In summary, therefore, the helical elements  10 A- 10 H of this preferred antenna are equally angularly spaced around the core  12  at intervals of 360°/N where N is the number of elements, and they are arranged in two groups each having N/2 elements that are of varying length owing to the varying distance of the rim  20 U of the sleeve  20  from the distal end face  12 D of the core  12 , which face is perpendicular to the central axis of the core. Each element executes substantially a half turn of the core in this embodiment, although alternative embodiments may employ elements having other integral multiples (1, 2, 3, . . . ) of a half turn or, indeed, may be quarter turn helices or multiples thereof. 
     The conductive sleeve  20 , the plating on the proximal end face  12 P of the core, and the outer shield  16  of the feeder together form a quarterwave balun that provides common-mode isolation of the radiating antenna element structure from the equipment to which the antenna is connected when installed when the antenna is operated at its operating frequency. Currents in the sleeve are, therefore, confined to the sleeve rim  20 U. Accordingly, at the operating frequency, the rim  20 U of the sleeve  20  and the helical elements of each pair  10 A,  10 E- 10 D,  10 H form a respective conductive loop connected to a balanced feed, currents travelling between the elements of each pair via the rim  20 U. 
     In this preferred embodiment of the invention, the circumference of the sleeve is equal to an integer number of guide wavelengths at the operating frequency. This has the effect of reinforcing the resonant mode arising from the resonance of the above-mentioned conductive loops formed by the pairs of helical elements and the rim at the operating frequency. In particular, as described in the above-mentioned British Patent Publication GB2346014A, the sleeve  20  acts as a resonant structure in itself, independently of the helical elements  10 A- 10 H. Thus, the rim  20 U of the sleeve, having an electrical length equal to the operating wavelength, is resonant in a ring mode. Reinforcement of the resonant mode due to the loops formed by the pairs of helical elements and the rim  20 U can be visualised by imagining a wave being injected onto the ring represented by the rim  20 U at the junction of each of the helical elements and the rim, the wave then travelling around the rim  20 U to form a spinning dipole, as described in GB2346014A. Owing to the electrical length of the rim  20 U, when the injected wave has traveled around the rim  20 U and arrives back at the injection point, the next wave is injected from the respective helical element, thereby reinforcing the first. This constructive combination of waves results from the resonant length of the rim. 
     Further details of the ring resonance and the action of the sleeve  20  and the plating on the proximal end surface  12 P of the core in contributing to the operation of the antenna with regard to circularly polarised electromagnetic waves are contained in the above-mentioned GB2346014A. Whilst the sleeve and plating of this embodiment of the invention are advantageous in that they provide both a balun function and a ring resonance, a ring resonance can also be provided independently by connecting the helical elements  10 A- 10 H to an annular conductor that encircles the core  12  and has both proximal and distal edges on the outer side surface portion of the core, rather than being in the form of a sleeve connected to the feeder shield conductor  16  to form an open-ended cavity, as in the present embodiment. Such a conductor may be comparatively narrow insofar as it may constitute an annular track the width of which is similar to the width of conductive tracks forming the helical elements  10 A- 10 H and, providing it has an electrical length corresponding to an integral multiple (1, 2, 3, . . . ) of the guide wavelength at the operating frequency, still produces a ring resonance reinforcing the resonant mode associated with the loops provided by the helical elements and their interconnection. An antenna having an annular track of this description is described hereinafter. 
     With regard to the resonant behaviour of the loops represented by the helical elements  10 A- 10 H and their interconnection, these combine such that at the operating frequency of the antenna, it operates in a mode of resonance in which the antenna is sensitive to circularly polarised signals. Each pair  10 AE,  10 BF,  10 CG,  10 DH of the helical elements has an associated resonance within a single operating frequency band of the antenna, and the pairs all co-operate to form a common circular polarisation resonance, as follows. The differing lengths of the antenna elements  10 A- 10 H result in 360°/N (45°) phase differences between currents in the different elements of each group  10 A- 10 D,  10 E- 10 H. In this resonant mode, currents flow around the rim  20 U between, on the one hand, the helical element of each pair  10 AE,  10 BF,  10 CG,  10 DH which is coupled to the inner feed conductor  18  and, on the other hand, that which is connected to the shield  16  by the coupling conductors of the laminate board  19 , as will be described below. The sleeve  20  and the plating on the proximal end face  12 P of the core together act as a trap preventing the flow of currents from the antenna elements  10 A- 10 H to the shield conductor  16  at the proximal end face  12 P of the core. 
     Operation of dielectrically loaded multifilar helical antennas having a balun sleeve is described in more detail in the above-mentioned British Patent Applications GB2292638A and GB2310543A. 
     The feeder transmission line performs functions other than simply as a line having a characteristic impedance of 50 ohms for conveying signals to or from the antenna element structure. Firstly, as described above, the shield  16  acts in combination with the sleeve  20  to provide common-mode isolation at the point of connection of the feed structure to the antenna element structure. The length of the shield conductor between (a) its connection with the plating  22  on the proximal end face  12 P of the core and (b) its connection to conductors on the laminate board  19 , together with the dimensions of the bore  12 B and the dielectric constant of the material filling the space between the shield  16  and the wall of the bore, are such that the electrical length of the shield  16  on its outer surface is, at least approximately, a quarter wavelength at the frequency of the required mode of resonance of the antenna, so that the combination of the conductive sleeve  20 , the plating  22  and the shield  16  promotes balanced currents at the connection of the feed structure to the antenna element structure. 
     In this preferred antenna, there is an insulative layer surrounding the shield  16  of the feed structure. This layer, which is of lower dielectric constant than the dielectric constant of the core  12 , diminishes the effect of the core  12  on the electrical length of the shield  16  and, therefore, on any longitudinal resonance associated with the outside of the shield  16 . Since the mode of resonance associated with the required operating frequency is characterised by voltage dipoles extending diametrically, i.e. transversely of the cylindrical core axis, the effect of the low dielectric constant sleeve on the required mode of resonance is relatively small due to the sleeve thickness being, at least in the preferred embodiment, considerably less than that of the core. It is, therefore, possible to cause the linear mode of resonance associated with the shield  16  to be de-coupled from the wanted mode of resonance. 
     The antenna has a main resonant frequency of 500 MHz or greater, the resonant frequency being determined by the effective electrical lengths of the helical antenna elements  10 A- 10 H and, to a lesser degree, by their width. The lengths of the elements, for a given frequency of resonance, are also dependent on the relative dielectric constant of the core material, the dimensions of the antenna being substantially reduced with respect to an air-cored quadrifilar antenna. 
     The antenna is especially suitable for satellite telephony and messaging in the Iridium band of from 1613.8 to 1626.5 MHz. In this case, the core  12  has a diameter of about 10 mm and the longitudinally extending antenna elements  10 A- 10 D have an average longitudinal extent (i.e. parallel to the central axis) of about 12 mm. The length of the conductive sleeve  20  is typically in the region of 5.5 mm. Precise dimensions of the antenna elements  10 A to  10 D can be determined in the design stage on a trial and error basis by undertaking eigenvalue delay measurements until the required phase differences are obtained. The diameter of the coaxial transmission line in the bore  12 B is in the region of 2 mm. 
     An alternative antenna having the features described above with reference to  FIGS. 1 to 4 , is resonant at 1575 MHz, the frequency of the L-band GPS service. In this case the core is 7.5 mm in diameter, the antenna elements have an average longitudinal extent of about 7 mm, and the balun sleeve length is about 2 mm. The relative dielectric constant of the core material is higher in this case, typically 76. 
     Further details of the feed structure will now be described. The feed structure comprises the combination of a coaxial 50 ohm line  16 ,  17 ,  18  and the planar laminate board  19  connected to a distal end of the line. The laminate board  19  is a multiple-layer printed circuit board (PCB) that lies flat against the distal end face  12 D of the core  12  in face-to-face contact. The largest dimension of the PCB  19  is smaller than the diameter of the core  12  so that the PCB  19  is fully within the periphery of the distal end face  12 D of the core  12 , as shown in  FIG. 1 . 
     In this embodiment, the PCB  19  is in the form of a disc centrally located on the distal face  12 D of the core. Its diameter is such that it overlies the arcuate inter-element coupling conductors  10 AD,  10 EH plated on the core distal face  12 D. As shown in  FIG. 6 , the PCB has a substantially central hole  32  which receives the inner conductor  18  of the coaxial feeder transmission line. Three off-centre holes  34  receive distal lugs  16 G of the shield  16 . Lugs  16 G are bent or “jogged” to assist in locating the PCB  19  with respect to the coaxial feeder structure. All four holes  32  are plated through. In addition, portions  19 P of the periphery of the PCB  19  are plated, the plating extending onto the proximal and distal faces of the board. 
     The PCB  19  is a multiple-layer board in that it has a plurality of insulative layers and a plurality of conductive layers. In this embodiment, the board has two insulative layers comprising a distal layer  36  and a proximal layer  38 . There are three conductor layers as follows: a distal layer  40 , an intermediate layer  42 , and a proximal layer  44 . The intermediate conductor layer  42  is sandwiched between the distal and proximal insulative layers  36 ,  38 , as shown in  FIG. 6 . Each conductor layer is etched with a respective conductor pattern, as shown in  FIGS. 7A to 7C . Where the conductor pattern extends to the peripheral portions  19 P of the PCB  19  and to the plated-through holes  32 ,  34 , the respective conductors in the different layers are interconnected by the edge plating and the hole plating respectively. As will be seen from the drawings showing the conductor patterns of the conductor layers  40 ,  42  and  44 , the intermediate layer  42  has a first conductor area  42 C in the shape of a fan or sector extending radially from a connection to the inner conductor  18  (when seated in hole  32 ) in the direction of the radial antenna element connection portions  10 AR- 10 DR. Directly beneath this conductive area  42 C, the proximal conductor layer  44  has a generally sector-shaped area  44 C extending from a connection with the shield  16  of the feeder (when received in plated via  34 ) to the board periphery  19 P overlying the arcuate or part-annular track  10 AD interconnecting the radial connection elements  10 AR- 10 DR. In this way, a shunt capacitor is formed between the inner feeder conductor  18  and the feeder shield  16 , the material of the proximal insulative layer  38  acting as the capacitor dielectric. This material typically has a dielectric constant greater than 5. 
     The conductor pattern of the intermediate conductive layer  42  is such that it has a second conductor area  42 L extending from the connection with the inner feeder conductor  18  to the second plated outer periphery  19 P so as to overlie the arcuate or part-annular track  10 EH. There is no corresponding underlying conductive area in the conductor layer  44 . The conductive area  42 L between the central hole  32  and the plated peripheral portion  19 P overlying the arcuate track  10 EH acts as a series inductance between the inner conductor  18  of the feeder and one of the groups of helical antenna elements  10 E- 10 H. 
     When the combination of the PCB  19  and the elongate feeder  16 - 18  is mounted to the core  12  with the proximal face of the PCB  19  in contact with the distal face  12 D of the core, aligned over the arcuate interconnection elements  10 AD and  10 EH as described above, connections are made between the peripheral portions  19 P and the underlying tracks on the core distal face  12 D to form a reactive matching circuit having a shunt capacitance and a series inductance. 
     The proximal insulative layer of the PCB  19  is formed of a ceramic-loaded plastics material to yield a relative dielectric constant for the layer  38  in the region of  10 . The distal insulative layer  36  can be made of the same material or one having a lower dielectric constant, e.g. FR-4 epoxy board. The thickness of the proximal layer  38  is much less than that of the distal layer  36 . Indeed, the distal layer  36  may act as a support for the proximal layer  38 . 
     Connections between the feeder line  16 - 18 , the PCB  19  and the conductive tracks on the distal face  12 D of the core are made by soldering or by bonding with conductive glue. The feeder  16 - 18  and the PCB  19  together form a unitary feeder structure when the distal end of the inner conductor  18  is soldered in the via  32  of the PCB  19 , and the shield lugs  16 G in the respective off-centre vias  34 . The feeder  16 - 18  and the PCB  19  together form a unitary feed structure with an integral matching network. 
     The shunt capacitance and the series inductance form a matching network between the coaxial transmission line at its distal end and the radiating antenna element structure of the antenna. The shunt capacitance and the series inductance together match the impedance presented by the coaxial line, physically embodied as shield  16 , insulative layer  17  and inner conductor  18 , when connected at its proximal end to radiofrequency circuitry having a 50 ohm termination, this coaxial line impedance being matched to the impedance of the antenna element structure at its operating frequency or frequencies. 
     As stated above, the feed structure is assembled as a unit before being inserted in the antenna core  12 , the laminate board  19  being fastened to the coaxial line  16 - 18 . Forming the feed structure as a single component, including the board  19  as an integral part, substantially reduces the assembly cost of the antenna, in that introduction of the feed structure can be performed in two movements: (i) sliding the unitary feed structure into the bore  12 B and (ii) fitting a conductive ferrule or washer around the exposed proximal end portion of the shield  16 . The ferrule may be a push fit on the shield component  16  or is crimped onto the shield. Prior to insertion of the feed structure in the core, solder paste is preferably applied to the connection portions of the antenna element structure on the distal end face  12 D of the core  12  and on the plating  22  immediately adjacent the respective ends of the bore  12 B. Therefore, after completion of steps (i) and (ii) above, the assembly can be passed through a solder reflow oven or can be subjected to alternative soldering processes such as laser soldering, inductive soldering or hot air soldering as a single soldering step. 
     Solder bridges formed between (a) conductors on the peripheral and the proximal surfaces of the board  19  and (b) the metallised conductors on the distal face  12 D of the core, and the shapes of the conductors themselves, are configured to provide balancing rotational meniscus forces during reflow soldering when the board is correctly orientated on the core. 
     The antenna described above has antenna elements which are plain helices of different physical lengths spaced uniformly around a cylindrical core. Variations are possible within the scope of the invention. These include an antenna in which the physical lengths of the helical elements are equal and differences in loop lengths are achieved instead by arranging for the conductive connection portions  10 AR- 10 HR plated on the distal end face  12 D of the core to be of different effective lengths. Alternatively, the widths of the helical elements  10 A- 10 H can be varied to yield different electrical lengths. It is also possible for the physical lengths of the helical elements and the electrical lengths of the loops represented by the laterally opposed pairs of helical elements and their interconnections also to be equal, the required response to circularly polarised electromagnetic waves and the associated radiation pattern being achieved solely as a result of the presence of a plurality of axially coextensive helical elements distributed around the core, especially with such elements being interconnected by an annular conductor such as sleeve  20  having an annular electrical length equal to the guide wavelength of the operating frequency, or a non-unity integral multiple thereof. One way of producing a variation in electrical length between successive members of each group  10 A- 10 D,  10 E- 10 H of helical elements is to arrange for the elements required to have a greater electrical length to be meandered. Indeed, all of the elements  10 E- 10 H may be meandered, but to different degrees. It is possible for one or both edges of certain elements to be meandered as well. One such antenna is illustrated in  FIGS. 8 and 9 . The respective “unrolled” conductive pattern of the outer cylindrical surface portion of the antenna is shown in  FIG. 10 . 
     Referring to  FIGS. 8 to 10 , this second antenna in accordance with the invention, like the first antenna described above with reference to  FIGS. 1 to 7 , has eight helical antenna elements in the form of conductive tracks plated on the outer cylindrical surface portion of the core  12 . As before, these helical elements are arranged in two groups  10 A- 10 D,  10 E- 10 H, the elements of each group being connected to respective radial tracks  10 AR- 10 DR,  10 ER- 10 HR and respective arcuate interconnecting tracks  10 AD,  10 EH, all plated on the distal face  12 D of the core  12 , as in the first embodiment. In this example, however, the helical elements of each group have meandered edges. As best seen in  FIG. 10 , the outer elements  10 A,  10 D;  10 E,  10 H of each group each have one edge meandered and the other edge as a plane helix, whereas the inner elements  10 B,  10 C;  10 F,  10 G of each group each have both edges meandered. The effect of such meandering is that the inner elements  10 B,  10 C;  10 F,  10 G have greater electrical lengths than the outer elements  10 A,  10 B;  10 E,  10 H. This arrangement is selected because, in this embodiment, the configuration of the radial connection tracks  10 AR- 10 HR and their interconnecting arcuate conductors  10 AD,  10 EH is such that the electrical lengths of the conductive paths between the feed structure  16 - 19  and the distal ends of the outer helical elements  10 A,  10 D;  10 E,  10 H of each group are greater than those of the corresponding connections to the upper ends of the inner helical elements  10 B,  10 C;  10 F,  10 G. The meandering, therefore, compensates for the differences in electrical lengths of the conductors on the distal face  12 D. In this example, the meandering is not used to effect differences in length between the helical elements  10 A- 10 H for the purpose of creating phase differences, although, as stated above, it is possible to use meandering for this purpose. In this embodiment, it is the non-planar sleeve rim  20 U that is used to effect a progression in helical element lengths, as in the first embodiment described above with reference to  FIGS. 1 to 7 . 
     The embodiments described so far are so-called “backfire” antennas inasmuch as they produce a radiation pattern directed outwardly from that end face of the antenna at which the radiating elements (the helical tracks and the respective connection conductors on the end face) are coupled to the feeder, the feeder comprising a transmission line passing through the core on the core axis. The invention is also applicable to an “endfire” antenna, the feed connection point of which is at the proximal end, i.e., the end opposite to that from which the antenna develops a maximum in the radiation pattern for circularly polarised waves. Such an antenna within the scope of the invention will now be described with reference to  FIGS. 11 and 12 . 
     Referring to  FIG. 11 , a third antenna in accordance with the invention has an antenna element structure with eight longitudinally extending helical antenna elements  10 A- 10 H, formed as plated metallic conductor tracks on the cylindrical outer surface portion of a cylindrical ceramic core  12 . An annular link conductor  20 C, positioned on the outer cylindrical surface of the core interconnects the antenna elements adjacent a distal end  12 D of the antenna. At the proximal end  12 P, eight radial connection elements  10 AR- 10 HR, formed as metallic tracks, are plated on the proximal end surface of the core. Each radial element  10 AR- 10 HR is electrically connected to a respective helical antenna element  10 A- 10 H and, as in the above-described backfire antennas, is connected to the other radial elements associated with the antenna elements of the same group  10 A- 10 D;  10 E- 10 H by an arcuate interconnecting conductor track  10 AD,  10 EH located between the axis of the core and the edge of the end surface upon which they are plated. 
     The annular link conductor  20 C acts as a common interconnecting conductor for the helical elements  10 A- 10 H at their distal ends. The proximal edge  20 CP of the linking conductor  20 C may be non-planar to cause the lengths of the helical elements  10 A- 10 H to vary in the same way that the sleeve rim  20 U of the first and second antennas described hereinabove is non-planar. As before, the lengths of the helical elements  10 A- 10 D,  10 E- 10 H in each group alter progressively according to their angular position at the core axis, the elements  10 A,  10 E being the longest in each group  10 A- 10 D,  10 E- 10 H, and the elements  10 D,  10 H being the shortest. The electrical length of the annular linking conductor  20 C equals the guide wavelength at the operating frequency of the antenna so that this linking conductor has a ring resonance as described above with reference to the sleeve rim  20 U of the first and second antennas (see  FIGS. 1 to 4  and  8  to  9 ). As in the case of the backfire antennas described above, other steps may be taken to vary the lengths of the helical elements  10 A- 10 H and their interconnecting conductors, as required. 
     The core  12  of the antenna shown in  FIG. 11  has no central passage but merely a circular recess  12 R in the proximal end face  12 P, centred on the core axis. This recess  12 R receives a central tab  50 T projecting from the distal edge of a multiple layer printed circuit board (PCB)  50  mounted to the proximal end face  12 P of the antenna core  12 , as shown most clearly in  FIG. 12 . Referring to  FIG. 12 , the PCB  50  has multiple electrically conductive layers and multiple insulative layers separating the electrically conductive layers, the conductive patterns of the layers being configured to form a balun  52  connected to the coupling nodes of the antenna formed by the arcuate conductor tracks  10 AD,  10 EH on the proximal end face  12 P of the antenna core  12 . The PCB carries a receiver front end circuit comprising a front end amplifier  54  housed within a screen  56  on one major face  52 A of the PCB  52 . 
     Connections between the balun  52  and the coupling nodes  10 AD,  10 EH of the antenna are made by four conductive brackets  58 , two on each major face  50 A,  50 B of the PCB  50 , adjacent the distal edge  50 D of the PCB  50 . In practice, the PCB  50  is secured to the antenna core  12  by a plastics collar, not shown in the drawings. 
     In this embodiment, the PCB  50  is centrally positioned with its major faces  50 A,  50 B parallel to the axis. The central plane of symmetry of the PCB  50 , parallel to and between the major faces  50 A,  50 B, bisects the proximal end face  12 P of the antenna core on a diameter perpendicular to a line on the end face  12 P passing between the two sets of radial elements  10 AR- 10 DR;  10 ER- 10 HR and their interconnecting arcuate conductors  10 AD,  10 EH so that the distal edge  50 D of the PCB  50  overlaps both arcuate interconnecting conductors  10 AD,  10 EH. The brackets  58  are located in registry with the respective interconnecting conductors  10 AD,  10 EH. Thus, each coupling node of the antenna is connected to the balun  52  by two respective connecting brackets  58 , one on each side of the PCB  50 . Each such pair of connecting brackets  58  is linked by a respective plated through-hole (via)  60  passing through the PCB  50  (see  FIG. 12 ). 
     The PCB  50  has five conductive layers  62 ,  63 ,  64 ,  65 ,  66  separated by four insulative layers  67 ,  68 ,  69 ,  70 . The middle or third conductive layer  64  is formed as a narrow conductive track extending along the axis of the antenna from one of the vias  60  interconnecting the conductive brackets  58  which are conductively bonded to the arcuate interconnecting conductor  10 EH, this elongate conductive track acting as the inner conductor of a shielded transmission line, the shield of which is formed by the two intermediate conductive layers  63 ,  65  which extend parallel to the track formed by the middle layer  64  and have vias (not shown) along their longitudinal edges to interconnect those edges along lines parallel to but spaced from the edges of the inner conductor  64 . The intermediate conductive layers  63 ,  65  forming the transmission line shield are connected to the conducting brackets  58  which overlie the arcuate interconnecting conductor  12 AD on the antenna distal end face  12 P, thereby connecting the transmission line as a feeder for the radiating antenna element structure of the antenna. 
     As will be seen from  FIG. 12 , the proximal ends of the transmission line conductors formed by layers  63 ,  64 ,  65  extend to the receiver circuitry and, specifically, has a via connection  72  to an input  74  of the amplifier  54 , the shield conductors  63 ,  65  being connected by another via  76  to the amplifier ground and the screen  56 . 
     The outer conductive layers  62 ,  66  of the PCB  50  are formed as conductive plates extending substantially the full width of the PCB and, thereby, overlapping the shield conductors formed by the intermediate layers  63 ,  65 . The distal edges of the plates formed by outer layers  62 ,  66 , i.e. the edges closest to the antenna, are open-circuit edges. In contrast, the proximal edges are interconnected by a line of vias  78  extending transversely of the PCB, these vias also connecting the proximal edges of the plates to the shield conductors formed by the intermediate layers  63 ,  65 . The relative dielectric constant of the insulative layers  67 ,  70  between the plates  62 ,  66  and the shield conductors  63 ,  65 , and the axial lengths of the plates  62 ,  66  are such that the electrical lengths of the plates  62 ,  66  in the axial direction are each a quarter guide wavelength at the operating frequency of the antenna. The PCB  50  thereby provides a balun matching the single-ended input of the receiver circuitry with the balanced feed connection of the antenna at the coupling nodes provided by the arcuate interconnecting conductors  10 AD,  10 EH. 
     This arrangement has several benefits. Firstly, the balun  52  chokes currents on the shield conductors formed by the intermediate layers  63 ,  65 , thereby preventing common-mode noise signals (generated, e.g., by other circuits in the equipment in which the antenna is mounted) flowing off the screen cage and entering the transmission line formed by the inner and intermediate layers  63 - 65 . In this manner the balun screens the transmission line from common-mode noise signals. The balun provides a balanced load for the antenna. Furthermore, the balun isolates the antenna such that only the antenna radiates. In addition, the resonant frequency of the system is determined by the antenna only, rather than the antenna together with exposed conductors of the link between the antenna and the receiving circuit. This means that the radiating and resonating conductor lengths are consistent. 
     As an alternative, the current choke can be formed by a half balun sleeve. In such an arrangement, only one balun plate is used. This has substantially the same effect as a full balun sleeve formed by two plates (layers  62 ,  66  in  FIG. 12 ). 
     One advantage of the octafilar antennas described above is that they may be placed closer to conductive structures without appreciable detuning. Referring to  FIG. 13 , a portable wireless terminal in accordance with one aspect of the invention has an antenna  100  located inside the terminal casing (not shown) and mounted adjacent an edge  102 E of a planar printed circuit board  102  which has an electrically conductive ground plane layer  102 G. The edge  102 E of the board  102  and that of the ground plane layer  102 G lies parallel to the outer cylindrical surface  12 C of the antenna core, upon which the helical elements  10 A- 10 H (see  FIG. 1 ) are plated and is spaced therefrom by a distance s which, in this embodiment, is in the region of 1 mm. The ground plane layer is spaced from the cylindrical surface  12 C by s over substantially the whole of the longitudinal or axial extent of the antenna structure formed by the helical elements. It will be noted that the ground plane layer  102 G lies in a plane containing the axis  100 A of the antenna or, at least, in a plane which is very close to and parallel to the axis  10 A. 
     The lack of detuning is thought to be due to the fact that, at the most sensitive parts of the helical antenna elements  10 A- 10 H ( FIG. 1 ), only one element out of the eight is affected by the proximity of the ground plane conductive layer  102 G. This represents a small percentage of the total complement of helical antenna elements compared with, for instance, one affected helical element of a quadrifilar helical antenna. 
     A further antenna in accordance with the invention has three pairs of helical antenna elements, as shown in  FIG. 14 . Referring to  FIG. 14 , in this hexafilar antenna the helical elements are, as in the case of the octafilar antenna described above, arranged in two groups, the elements of each group being connected to a respective coupling node. Thus, three coextensive helical conductive tracks  10 A,  10 B,  10 C plated or otherwise metallised on the cylindrical outer surface of the cylindrical core  12  are coupled to one side of the feeder via radial tracks  10 AR,  10 BR,  10 CR, and a further three helical tracks  10 E,  10 F,  10 G are coupled to the other side of the feeder by respective radial tracks  10 ER,  10 FR,  10 GR. As before, two arcuate conductors  10 AC,  10 EG plated on the core distal face  12 D adjacent the end of the board  12 B interconnect the respective radial tracks in order that coupling to the feeder shield and inner conductors may be achieved via the laminate board  19  described above. 
     The helical tracks  10 A- 10 C,  10 E- 10 G are uniformly angularly spaced around the axis at 120° intervals, a combined circular polarisation resonance being achieved in a manner similar to that described above in connection with the octafilar antenna by varying the individual electrical lengths of the loops formed by the paired elements  10 A,  10 E;  10 B,  10 F;  10 C,  10 G. It has been found that the physical lengths of the helical elements  10 A- 10 C,  10 E  10 G are advantageously shorter in the case of the outer elements  10 A,  10 C,  10 E,  10 G of the two groups than the middle or inner conductor tracks  10 B,  10 F. A suitable profile for the balun rim  20 U is shown diagrammatically in  FIG. 5 . It will be appreciated that, in this diagram, the magnitude of the height variations of the rim  20 U is greatly magnified for clarity of illustration of the principle. 
     A particular property of a hexafilar antenna such as that described above with reference to  FIGS. 14 and 15  is that its bandwidth is greater than the bandwidth of comparable quadrifilar and octafilar antennas. This is because the pair of inner helical tracks may be regarded as a bifilar loop, the hexafilar antenna constituting the combination of a quadrifilar antenna and a bifilar antenna. The resonant bandwidth of a bifilar loop is greater than that of the combination of two loops in a quadrifilar arrangement, because the resonance of a quadrifilar arrangement, for circular polarisation, depends on a particular phase relationship that exists only over a narrow band of frequencies. 
     The bifilar resonance couples with those of the outer tracks so as to broaden the combined circular polarisation resonance. This is shown in the graph of  FIG. 16  which is a plot of the amplitudes of individual voltages on the helical conductor tracks  10 A- 10 C,  10 E- 10 G with respect to frequency. These plots are obtained by capacitive probes mounted close to the junctions of the respective tracks with the balun rim  20 U, in a manner similar to that described in our U.S. Pat. No. 6,886,237. It will be noted that the inner tracks  10 B,  10 F exhibit a broad resonance and, in particular, exhibit a dip or “saddle” in the region R of the intersection of the responses of the outer elements  10 A,  10 C,  10 E,  10 G, as shown in  FIG. 16 . This is evidence of the sharing of energy, i.e. coupling, between the elements at the frequency of operation of the antenna. 
     With the antenna described and shown, a 3 dB fractional bandwidth of at least 1% can be expected, with a figure of 1.2%.