Patent Publication Number: US-2011051790-A1

Title: Radio communication device and method

Description:
CROSS-REFERENCE TO RELATED APPLICATION(S) 
     This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2009-202698, filed on Sep. 2, 2009, the entire contents of which are incorporated herein by reference. 
     FIELD 
     Embodiments of the invention relate to a radio communication device and method. 
     BACKGROUND 
     To reduce the size and cost of radio communication devices, devices employing a direct conversion system have been increasing in recent years. According to the direct conversion system, a transmitter directly up-converts I- and Q-channel baseband signals into a transmission carrier frequency, and a receiver directly down-converts a received signal into I- and Q-channel baseband signals. 
     The direct conversion system does not require an intermediate filter and image rejection in an IF (Intermediate Frequency), and is expected to result in a reduction in size and cost. However, DC (Direct Current) offset, frequency offset, phase noise, IQ imbalance, and so forth occur as phenomena in an RF (Radio Frequency) unit of a radio communication device. These phenomena deteriorate communication characteristics. 
     A variety of methods have been studied to compensate for these imperfections of the radio unit (RF unit). A major one of the methods performs channel estimation with the use of a preamble (training signal) included in a received signal, to thereby correct the IQ imbalance, the frequency offset, and the DC offset. If the difference in amplitude and phase between the I and Q channels varies by frequency, however, the variation manifests as the deterioration of the flatness of the signal band. Further, if the variation is substantial, it is difficult to perform the compensation based on the channel estimation. 
     In view of the above, there is a method for compensating for the IQ imbalance, which provides beforehand a correction factor to a transmission signal to compensate for the IQ imbalance. For example, a method has been known which compensates for the gain imbalance and the phase shift occurring in baseband filters provided for the I- and Q-channels in a transmitter (Japanese Laid-open Patent Publication No. 2006-523057, for example). 
     SUMMARY 
     According to an aspect of the invention, a radio communication device includes a first filter configured to receive an input of a first transmission signal, a second filter configured to receive an input of a second transmission signal orthogonal to the first transmission signal, a radio unit configured to perform quadrature modulation on signals output from the first filter and the second filter, and produce a radio signal, a switch configured to provide, when a first test signal and a second test signal are present, the radio signal to a reception unit as a corresponding test radio signal, and a baseband signal processing unit configured to compensate for in-phase/quadrature imbalance by outputting the first test signal to the first filter, output the second test signal to the second filter, and calculate, on a basis of the test radio signal received via the reception unit, a correction factor to be applied to the first transmission signal and the second transmission signal. 
     The object and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the claims. 
     It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention, as claimed. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of an exemplary radio communication device according to a first embodiment; 
         FIG. 2  is a block diagram of an exemplary radio communication device according to a second embodiment; 
         FIG. 3  is a diagram illustrating quadrature modulation by an IQ modulation unit; 
         FIG. 4  is a diagram illustrating a spectrum obtained when test signals are normally quadrature-modulated; 
         FIG. 5  is a diagram illustrating a spectrum obtained when test signals are not normally quadrature-modulated; 
         FIGS. 6A to 6D  are diagrams illustrating spectra of respective sections of the radio communication device illustrated in  FIG. 2 ; 
         FIG. 7  is a diagram illustrating a data configuration example of a correction factor table; 
         FIG. 8  is a block diagram of an exemplary transmission signal generation unit in  FIG. 2 ; 
         FIG. 9  is a block diagram of an exemplary radio communication device according to a third embodiment; and 
         FIG. 10  is a block diagram of an exemplary radio communication device according to a fourth embodiment. 
     
    
    
     DESCRIPTION OF THE EMBODIMENTS 
     The existing method of compensating for the IQ imbalance, however, compensates for the IQ imbalance of the baseband filter. Therefore, the method has an issue of lack of compensation for the IQ imbalance of the radio unit at a subsequent stage of the baseband filter. 
     The present case has been made in view of the above-described circumstances, and it is an object of the invention to provide a radio communication device capable of compensating for the IQ imbalance of the filter and the radio unit. 
     To solve the above-described issue, a radio communication device and method which performs radio communication is provided. The radio communication device includes a first filter, a second filter, a radio unit, a switch, and a baseband signal processing unit. The first filter is configured to receive an input of a first transmission signal. The second filter is configured to receive an input of a second transmission signal orthogonal to the first transmission signal. The radio unit is configured to perform quadrature modulation on the signals output from the first filter and the second filter, and output a radio signal. The switch is configured to switch, when a first test signal and a second test signal are present, a test radio signal (output from the radio unit) to a reception unit which receives a radio received signal. The baseband signal processing unit is configured to output the first test signal to the first filter, output the second test signal to the second filter, and calculate, on the basis of the test radio signal output from the reception unit, a correction factor to be applied to the first transmission signal and the second transmission signal to compensate for IQ imbalance occurring in the first filter, the second filter, and the radio unit. 
     The disclosed radio communication device is capable of compensating for the IQ imbalance of a filter and a radio unit. 
     A first embodiment will be described in detail with reference to the drawings. 
       FIG. 1  is a block diagram of a radio communication device according to the first embodiment. As illustrated in  FIG. 1 , the radio communication device includes a transmission signal generation unit  1 , a test signal generation unit  2 , switches  3  and  6 , a first filter  4   a,  a second filter  4   b,  a radio unit  5 , a reception unit  7 , and a correction factor calculation unit  8 . 
     The transmission signal generation unit  1  generates a first transmission signal and a second transmission signal, orthogonal to the first transmission signal, which are to be transmitted to another communication party. The transmission signal generation unit  1  applies a correction factor calculated by the correction factor calculation unit  8  to the first transmission signal and the second transmission signal, and outputs resultant signals to the switch  3 . 
     The test signal generation unit  2  generates a first test signal and a second test signal. The switch  3  outputs one of the first transmission signal and the first test signal to the first filter  4   a,  and outputs one of the second transmission signal and the second test signal to the second filter  4   b.    
     The first filter  4   a  receives an input of the first transmission signal or the first test signal. The second filter  4   b  receives an input of the second transmission signal or the second test signal. Each of the first filter  4   a  and the second filter  4   b  performs band limitation on the signal input thereto, and outputs a resultant signal to the radio unit  5 . 
     The radio unit  5  performs quadrature modulation on the signals output from the first filter  4   a  and the second filter  4   b,  and outputs a radio signal. 
     When the first transmission signal and the second transmission signal are output from the switch  3  and a radio transmission signal is output from the radio unit  5 , the switch  6  switches connections such that the radio transmission signal is output to an antenna. Further, when a radio received signal is received from the other communication party, the switch  6  switches connections such that the radio received signal received by the antenna is output to the reception unit  7 . Further, when the first test signal and the second test signal are output from the switch  3  and a test radio signal is output from the radio unit  5 , the switch  6  switches connections such that the test radio signal is returned to the reception unit  7 . 
     The reception unit  7  performs the processing of receiving an input signal. For example, the reception unit  7  performs down-conversion on an input signal. 
     On the basis of the test radio signal output from the reception unit  7 , the correction factor calculation unit  8  calculates the correction factor for compensating for the IQ imbalance occurring in the first filter  4   a,  the second filter  4   b,  and the radio unit  5 . As described above, the correction factor is applied to the first transmission signal and the second transmission signal. Thereby, the IQ imbalance of the first filter  4   a,  the second filter  4   b,  and the radio unit  5  is compensated. 
     The radio communication device is thus configured to output the first test signal and the second test signal to the first filter  4   a  and the second filter  4   b,  respectively, return the test signals to the reception unit  7  via the radio unit  5 , and calculate the correction factor. Accordingly, it is possible to compensate for the IQ imbalance of the first filter  4   a,  the second filter  4   b,  and the radio unit  5 . 
     Subsequently, a second embodiment will be described.  FIG. 2  is a block diagram of a radio communication device according to the second embodiment. As illustrated in  FIG. 2 , the radio communication device includes a baseband signal processing unit  11 , DACs (Digital to Analog Converters)  12   a  and  12   b,  LPFs (Low Pass Filters)  13   a,    13   b,    22   a,  and  22   b,  an IQ modulation unit  14 , a PA (Power Amplifier)  15 , switches  16 ,  17 ,  20 , and  26 , an ATT (ATTenuater)  18 , an LNA (Low Noise Amplifier)  19 , an IQ demodulation unit  21 , ADCs (Analog to Digital Converters)  23   a  and  23   b,  a local oscillator  24 , and a frequency shifter  25 . The radio communication device is applied to, for example, a mobile phone and a radio base station. The radio communication device performs, for example, radio communication according to the OFDM (Orthogonal Frequency Division Multiplexing) system. Further, the baseband signal processing unit  11  may be realized by a baseband processing LSI (Large Scale Integration). 
     The radio unit  5  of  FIG. 1  may include the IQ modulation unit  14  and the PA (Power Amplifier)  15  of  FIG. 2 . Further, the reception unit  7  of  FIG. 1  may include the LNA (Low Noise Amplifier)  19  and the IQ demodulation unit  21 . 
     The DACs  12   a  and  12   b,  the LPFs  13   a  and  13   b,  the IQ modulation unit  14 , the PA  15 , and the switch  16  form a transmission unit. The LNA  19 , the switch  20 , the IQ demodulation unit  21 , the LPFs  22   a  and  22   b,  and the ADCs  23   a  and  23   b  form a reception unit. The IQ modulation unit  14  and the PA  15  form an RF unit of the transmission unit. The LNA  19  and the IQ demodulation unit  21  form an RF unit of the reception unit. The local oscillator  24  and the frequency shifter  25  form an RF unit shared by the transmission unit and the reception unit. 
     The baseband signal processing unit  11  generates test signals for calculating a correction factor for compensating for the IQ imbalance of the LPFs  13   a  and  13   b  and the RF unit of the transmission unit. In accordance with the switching of the switches  16  and  20 , the baseband signal processing unit  11  receives the generated test signals through the device without radio-transmitting the test signals. Then, on the basis of the received test signals, the baseband signal processing unit  11  calculates the correction factor for compensating for the IQ imbalance. 
     The baseband signal processing unit  11  generates I- and Q-channel digital baseband signals to be transmitted to the other communication party. The baseband signal processing unit  11  applies the above-described correction factor to the generated baseband signals to compensate for the IQ imbalance of the LPFs  13   a  and  13   b  and the RF unit of the transmission unit. 
     The DACs  12   a  and  12   b  convert the baseband signals output from the baseband signal processing unit  11  into analog signals. The LPFs  13   a  and  13   b  cut off high-frequency components of the baseband signals converted into the analog signals, and allow low-frequency components of the baseband signals to pass therethrough. 
     The IQ modulation unit  14  performs quadrature modulation on the analog baseband signals output from the LPFs  13   a  and  13   b,  and directly up-converts the baseband signals into a radio frequency (RF). 
     The IQ modulation unit  14  includes multipliers  41   a  and  41   b  and a quadrature phase generator (0°/90° in  FIG. 2 )  42 . The quadrature phase generator  42  receives an input of an RF local signal output from the local oscillator  24 . The quadrature phase generator  42  sets the phase of the local signal to 0° and 90°, and outputs resultant signals to the multipliers  41   a  and  41   b.    
     The multiplier  41   a  multiplies the I-channel baseband signal output from the LPF  13   a  by the 0° phase local signal, to thereby directly convert the I-channel baseband signal into the RF. The multiplier  41   b  multiplies the Q-channel baseband signal output from the LPF  13   b  by the 90° phase local signal, to thereby directly convert the Q-channel baseband signal into the RF. The RF-converted I- and Q-channel baseband signals (radio signals) are synthesized and output to the PA  15 . 
     The PA  15  amplifies the radio signal output from the IQ modulation unit  14 . The switch  16  outputs the radio signal output from the PA  15  to one of the switch  17  and the ATT  18 . When the baseband signals to be transmitted to the other communication party (transmission signals) are output from the baseband signal processing unit  11 , the switch  16  switches outputs such that a radio transmission signal output from the PA  15  is radio-transmitted via an antenna. When the test signals are output from the baseband signal processing unit  11 , the switch  16  switches outputs such that the test radio signal output from the PA  15  is received by the baseband signal processing unit  11  via the ATT  18  and the reception unit. The ATT  18  attenuates the test radio signal output from the switch  16 . 
     The switch  17  switches between the connection of the output of the switch  16  with the antenna and the connection of the antenna with the input of the LNA  19 . When a transmission signal is radio-transmitted to the other communication party, the switch  17  performs switching such that the output of the switch  16  and the antenna are connected to each other. When a radio received signal is received from the other communication party, the switch  17  performs switching such that the antenna and the input of the LNA  19  are connected to each other. 
     The LNA  19  amplifies the radio received signal received by the antenna. The switch  20  outputs, to the IQ demodulation unit  21 , one of the radio received signal output from the LNA  19  and the test radio signal output from the ATT  18 . When the test signals are output from the baseband signal processing unit  11 , the switch  20  performs switching such that the test radio signal output from the ATT  18  is output to the IQ demodulation unit  21 . When the radio received signal is received from the other communication party, the switch  20  performs switching such that the radio received signal received by the antenna is output to the IQ demodulation unit  21 . 
     When the radio received signal received from the other communication party is output from the switch  20 , the IQ demodulation unit  21  performs quadrature demodulation on the radio received signal, and directly down-converts the radio received signal into the frequency of the baseband signals. When the test radio signal is output from the ATT  18 , the IQ demodulation unit  21  down-converts the test radio signal into the IF. 
     The IQ demodulation unit  21  includes multipliers  51   a  and  51   b  and a quadrature phase generator  52 . The quadrature phase generator  52  receives an input of the RF local signal output from the local oscillator  24 . Further, the quadrature phase generator  52  receives an input of a local signal frequency-shifted by the frequency shifter  25  to a lower frequency than the RF (IF-shifted signal). When the radio received signal is received from the other communication party, the quadrature phase generator  52  receives an input of the local signal of the local oscillator  24 . When the test signals are output from the baseband signal processing unit  11 , the quadrature phase generator  52  receives an input of the IF-shifted signal. The quadrature phase generator  52  sets the respective phases of the local signal and the IF-shifted signal to 0° and 90°, and outputs resultant signals to the multipliers  51   a  and  51   b.    
     The multiplier  51   a  multiplies the radio received signal output from the switch  20  by the 0° phase local signal, and outputs an I-channel baseband signal. The multiplier  51   b  multiplies the radio received signal output from the switch  20  by the 90° phase local signal, and outputs a Q-channel baseband signal. Further, the multiplier  51   a  multiplies the test radio signal output from the switch  20  by the IF-shifted signal to convert the test radio signal into the IF, and outputs a resultant signal. The test radio signal has the frequency thereof down-converted into the IF, but is not subjected to quadrature demodulation. 
     The LPFs  22   a  and  22   b  cut off high-frequency components of the signals output from the IQ demodulation unit  21 , and allow low-frequency components of the signals to pass therethrough. The ADCs  23   a  and  23   b  convert the analog signals output from the LPFs  22   a  and  22   b  into digital signals, and output the digital signals to the baseband signal processing unit  11 . 
     The local oscillator  24  outputs the RF local signal. The frequency shifter  25  frequency-shifts the RF of the local signal output from the local oscillator  24  to a lower frequency, and outputs the IF-shifted signal. When the test signals are output from the baseband signal processing unit  11 , the frequency shifter  25  outputs the IF-shifted signal. 
     When the transmission signals are output from the baseband signal processing unit  11 , the switch  26  switches connections such that a short circuit is caused between the input and output of the frequency shifter  25  to output the local signal of the local oscillator  24  to the IQ demodulation unit  21 . 
     The baseband signal processing unit  11  will be described in detail. The baseband signal processing unit  11  includes a transmission signal generation unit  31 , a correction factor table  32 , a test signal generation unit  33 , a switch  34 , a received signal processing unit  35 , a DDC (Digital Down Converter)  36 , an FFT (Fast Fourier Transform unit)  37 , a correction factor calculation unit  38 , and a frequency control unit  39 . 
     The transmission signal generation unit  31  places, on the frequency axis, transmission data to be transmitted to the other communication party, performs mapping (subcarrier modulation) of the transmission data onto the QPSK (Quadrature Phase Shift Keying) or 16QAM (Quadrature Amplitude Modulation) constellation, and thereafter performs IFFT (Inverse FFT) processing on the transmission data. Then, the transmission signal generation unit  31  adds guard intervals to the IFFT-processed signals, to thereby generate I- and Q-channel digital baseband signals. 
     The correction factor table  32  stores the correction factor for compensating for the IQ imbalance of the LPFs  13   a  and  13   b  and the RF unit of the transmission unit. The transmission signal generation unit  31  applies the correction factor to the signals subjected to the subcarrier modulation, to thereby compensate for the IQ imbalance of the LPFs  13   a  and  13   b  and the RF unit of the transmission unit. 
     The test signal generation unit  33  generates the test signals for calculating the correction factor for the IQ imbalance. The test signal generation unit  33  generates the digital test signals such that the analog test signals output from the DACs  12   a  and  12   b  have sine waves different in phase from each other by 90°. 
     The switch  34  switches the outputs of the baseband signals output from the transmission signal generation unit  31  and the test signals output from the test signal generation unit  33 . When the correction factor is calculated, the switch  34  performs switching such that the test signals output from the test signal generation unit  33  are output to the DACs  12   a  and  12   b.  When the transmission signals are transmitted to the other communication party, the switch  34  performs switching such that the baseband signals output from the transmission signal generation unit  31  are output to the DACs  12   a  and  12   b.  The calculation of the correction factor is performed, for example, upon power-on of the radio communication device or periodically. The periodical calculation of the correction factor may be performed when the transmission signals are not transmitted to the other communication party. 
     The received signal processing unit  35  performs, for example, decoding processing of the received signals digitally converted by the ADCs  23   a  and  23   b,  to thereby obtain received data transmitted by the other communication party. 
     The DDC  36  performs digital down-conversion on the IF test radio signal digitally converted by the ADC  23   a,  to thereby perform digital quadrature demodulation on the test radio signal. The DDC  36  may also perform digital down-conversion on the IF test radio signal digitally converted by the ADC  23   b.    
     The FFT  37  performs Fourier transform on the I- and Q-channel test radio signals subjected to the digital down-conversion by the DDC  36 . The correction factor calculation unit  38  calculates the correction factor on the basis of the spectrum of the test radio signals subjected to the Fourier transform, and stores the correction factor in the correction factor table  32 . The frequency control unit  39  controls the frequency of the local signal of the local oscillator  24 . 
     The generation of the test signals and the IQ imbalance will be described. To generate the test signals and calculate the correction factor, the switch  16  is first switched to connect the output of the PA  15  to the ATT  18 . Further, the switch  20  is switched to connect the ATT  18  to the IQ demodulation unit  21 . Thereby, the test signals output from the baseband signal processing unit  11  are returned to the reception unit without being radio-transmitted, and are input to the baseband signal processing unit  11 . Further, the switch  26  is brought into the open state such that the local signal of the local oscillator  24  is frequency-shifted by the frequency shifter  25  and output to the IQ demodulation unit  21 . 
     In the OFDM system, if imbalance in amplitude and phase occurs between the I and Q channels, the orthogonality fluctuates, and communication characteristics are deteriorated. In view of this, the test signal generation unit  33  may generate the test signals separately from the transmission signals to be transmitted to the other communication party, and the correction factor calculation unit  38  calculates the correction factor for compensating for the IQ imbalance on the basis of the test radio signals transmitted through the device. 
     The I- and Q-channel test signals output from the DACs  12   a  and  12   b  are represented by the following equations (1) and (2). 
         X   testI ( t )=cos ω l   t    (1)
 
         X   testQ ( t )=−sin ω l   t    (2)
 
     Herein, the equation ω l =2πf l  holds, wherein f l  represents the frequency of the test signal, and l represents the subcarrier number. The frequency f l  is prepared for each subcarrier used for data transmission. 
     If it is difficult to prepare the test signal for all subcarriers due to, for example, the limitation of the processing time, the test signal may be prepared for some of the subcarriers. In this case, the test signal is prepared to be dispersed across the subcarriers. 
     The test signals of the above equations (1) and (2) are subjected to quadrature modulation by the IQ modulation unit  14 . 
       FIG. 3  is a diagram illustrating quadrature modulation by the IQ modulation unit  14 .  FIG. 3  illustrates the multipliers  41   a  and  41   b  of the IQ modulation unit  14  illustrated in  FIG. 2 .  FIG. 3  further illustrates an adder  61  not illustrated in  FIG. 2 . In  FIG. 3 , the illustration of the quadrature phase generator  42  is omitted. 
     The quadrature phase generator  42  (illustrated in  FIG. 2 ) receives an input of the local signal output from the local oscillator  24 . The quadrature phase generator  42  outputs the local signals represented by the following equations (3) and (4), which are different in phase from each other by 90°, to the multipliers  41   a  and  41   b,  respectively. 
       L I =cos ω c t   (3)
 
         L   Q =−sin ω c   t    (4)
 
     Herein, ω c  represents a carrier frequency (RF). 
     The multiplier  41   a  multiplies the test signal represented by the equation (1) by the local signal represented by the equation (3). The multiplier  41   b  multiplies the test signal represented by the equation (2) by the local signal represented by the equation (4). The adder  61  adds up the signals output from the multipliers  41   a  and  41   b,  and outputs a signal x(t). Therefore, the signal x(t) output from the IQ modulation unit  14  is represented by the following equation (5). 
         x ( t )=cos(ω l   t )cos(ω c   t )−sin(ω l   t )sin(ω c   t )=(½)cos(ω l +ω c ) t    (5)
 
     According to the equation (5), the frequency of the signal output from the IQ modulation unit  14  is represented as ω l +ω c , and the frequency of the test radio signal is shifted from the carrier frequency ω c  to a higher frequency by ω l . 
     Further, the equation (5) is resolved and expressed in the following equations (6) and (7). 
       cos(ω l   t )cos(ω c   t )=(½){ cos(ω c +ω l ) t +cos(ω c −ω l ) t}   (6)
 
       sin(ω l   t )sin(ω c   t )=(½){ cos(ω c +ω l ) t −cos(ω c −ω l ) t}   (7)
 
       FIG. 4  is a diagram illustrating a spectrum obtained when the test signals are normally quadrature-modulated. According to the equations (6) and (7), if the I- and Q-channel test signals are quadrature-modulated with the 90° phase difference therebetween accurately maintained, the signal shifted from the carrier frequency ω c  to a lower frequency by ω l  is canceled. As illustrated in  FIG. 4 , therefore, the spectrum of the test radio signal output from the IQ modulation unit  14  remains only in a high-frequency region. 
       FIG. 5  is a diagram illustrating a spectrum obtained when the test signals are not normally quadrature-modulated. According to the equations (6) and (7), if the I- and Q-channel test signals are not quadrature-modulated with the 90° phase difference therebetween accurately maintained, the signal shifted to a lower frequency is not canceled. As illustrated in  FIG. 5 , therefore, the spectrum of the test radio signal output from the IQ modulation unit  14  also remains in a low-frequency region. Consequently, the remaining spectrum causes noise and deteriorates communication characteristics. 
       FIGS. 6A to 6D  are diagrams illustrating spectra of respective sections of the radio communication device illustrated in  FIG. 2 .  FIG. 6A  illustrates the spectrum of the test radio signal in the output from the PA  15  in  FIG. 2 .  FIG. 6B  illustrates the spectrum of the test radio signal in the output from the multiplier  51   a  of the IQ demodulation unit  21 .  FIG. 6C  illustrates the spectrum of the test radio signal in the output from the LPF  22   a.    FIG. 6D  illustrates the spectrum of the test radio signal in the output from the DDC  36 . 
     It is now assumed that the IQ imbalance occurs in the LPF  13   a  or  13   b,  the IQ modulation unit  14 , or the PA  15 . In this case, the spectrum appears in a frequency region lower than the carrier frequency ω c  in the output from the PA  15 , as illustrated in  FIG. 6A . 
     The test radio signal output from the PA  15  is output to the IQ demodulation unit  21  by the switches  16  and  20 . The test radio signal is multiplied by the IF-shifted signal output from the frequency shifter  25  by the multiplier  51   a  of the IQ demodulation unit  21 . 
     The test radio signal input to the multiplier  51   a  is down-converted into the IF by the IF-shifted signal, and the test radio signal in the output from the multiplier  51   a  has a spectrum as illustrated in  FIG. 6B . Herein, the frequency of the IF-shifted signal is represented as ω L0 (ω L0 &lt;ω c ). A frequency ω IF  of the IF has the relationship represented by the following equation (8). 
       ω IF =ω c −ω L0    (8)
 
     Due to the down-conversion into the IF, the spectrum also appears in a region corresponding to the equation ω=ω c +ω L0 . 
     The LPF  22   a  cuts off high frequencies of the test radio signal down-converted into the IF and output from the multiplier  51   a.  As illustrated in  FIG. 6C , therefore, the spectrum of the test radio signal in the output from the LPF  22   a  is cut off in an ω region and remains in an ω IF  region. 
     The test radio signal output from the LPF  22   a  is digitally converted by the ADC  23   a  and input to the DDC  36 . The DDC  36  multiplies the digital test radio signal output from the ADC  23   a  by the following equation (9), to thereby perform digital down-conversion on the test radio signal. 
         y ( t )= e   jω     IF     t    (9)
 
     With the multiplication using the equation (9), the spectrum of the digitally demodulated test signal is obtained from the DDC  36 , as illustrated in  FIG. 6D . 
     The calculation of the correction factor will be described. The test signal digitally demodulated by the DDC  36  is subjected to spectrum calculation by the FFT  37 . The correction factor calculation unit  38  retrieves the maximum value of the spectrum calculated by the FFT  37 . The correction factor calculation unit  38  calculates the ratio between the spectrum having the retrieved maximum value (the frequency of the transmitted test signal) and a negative spectrum paired with the spectrum having the maximum value, i.e., the DU (Desired to Undesired signal) ratio. That is, the correction factor calculation unit  38  calculates the DU ratio between the upper sideband and the lower sideband illustrated in  FIG. 6D . 
     The test signal generation unit  33  outputs the test signals while changing the amplitude and phase of the I- and Q-channel test signals represented by the equations (1) and (2). The correction factor calculation unit  38  calculates the DU ratio for each of the test signals, the amplitude and phase of which are changed. The correction factor calculation unit  38  stores, in the correction factor table  32 , the amplitude ratio of the present amplitude to the initial amplitude value and the phase difference of the present phase from the initial phase value obtained when the DU ratio falls to or below a predetermined threshold value, e.g., 25 dB. 
     For example, it is now assumed that the test signal generation unit  33  outputs a test signal of Ae j θ, wherein A and θ represent the amplitude and the phase, respectively. The above-described expression of the amplitude-phase representation is denoted by complex notation I+jQ. 
     The test signal generation unit  33  generates, as the test signal having the initial values, a test signal having values of A=1 and θ=0, for example. The test signal generation unit  33  outputs the test signal while changing the values of A and θ. Herein, it is assumed that the present amplitude and phase obtained when the DU ratio falls to or below a threshold value are represented as A=A p  and θ=θ p , respectively. Then, an amplitude ratio A p /A and a phase difference θ p  are stored in the correction factor table  32 . The test signal is generated for all subcarriers or predetermined selected ones of the subcarriers, and the amplitude ratio and the phase difference are calculated while the amplitude and phase of the test signal are changed. 
     A method of changing the amplitude and phase of the test signal will be described. It is now assumed that the test signal in Subcarrier No. l generated by the test signal generation unit  33  has an amplitude A l  and a phase θ l , wherein l represents the subcarrier number. Herein, there is a method of calculating the DU ratio equal to or less than a predetermined threshold value by retrieving all values of A l  (0&lt;A l &lt;a, wherein a represents a positive real number) and θ l  (−π&lt;θ l &lt;π). In the following, however, description will be made of the steepest descent method of simultaneously retrieving a plurality of parameters. 
     The steepest descent method changes (updates) the amplitude and phase of the test signal on the basis of the following equation (10). 
     
       
         
           
             
               
                 
                   
                     ( 
                     
                       
                         
                           
                             A 
                             l 
                             
                               ( 
                               
                                 k 
                                 + 
                                 1 
                               
                               ) 
                             
                           
                         
                       
                       
                         
                           
                             θ 
                             l 
                             
                               ( 
                               
                                 k 
                                 + 
                                 1 
                               
                               ) 
                             
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     
                       ( 
                       
                         
                           
                             
                               A 
                               l 
                               
                                 ( 
                                 k 
                                 ) 
                               
                             
                           
                         
                         
                           
                             
                               θ 
                               l 
                               
                                 ( 
                                 k 
                                 ) 
                               
                             
                           
                         
                       
                       ) 
                     
                     - 
                     
                       α 
                        
                       
                         ( 
                         
                           
                             
                               
                                 
                                   ∂ 
                                   
                                     D 
                                     
                                       ( 
                                       k 
                                       ) 
                                     
                                   
                                 
                                 / 
                                 
                                   ∂ 
                                   
                                     A 
                                     l 
                                     
                                       ( 
                                       k 
                                       ) 
                                     
                                   
                                 
                               
                             
                           
                           
                             
                               
                                 
                                   ∂ 
                                   
                                     D 
                                     
                                       ( 
                                       k 
                                       ) 
                                     
                                   
                                 
                                 / 
                                 
                                   ∂ 
                                   
                                     θ 
                                     l 
                                     
                                       ( 
                                       k 
                                       ) 
                                     
                                   
                                 
                               
                             
                           
                         
                         ) 
                       
                     
                   
                 
               
               
                 
                   ( 
                   10 
                   ) 
                 
               
             
           
         
       
     
     Herein, α represents the value determining the update rate, and is a positive real number. Further, A l   (k)  and θ l   (k)  represent the respective values of the amplitude and phase obtained by the k times of updates. Further, D (k)  represents the DU ratio obtained by the k-th update. For example, values of A l   (0) =1 and θ l   (0) =0 are set as the initial values, and the DU ratio is measured while the values of A l  and θ l  are changed by minute amounts. Then, the obtained results are substituted in the equation (10) to simultaneously update the amplitude and phase. 
     The calculation is repeated until the DU ratio falls to or below the preset threshold value. This calculation is performed in all subcarriers used for data communication or predetermined selected ones of the subcarriers. The amplitude ratio and the phase difference obtained for each of the subcarriers are stored in the correction factor table  32 . 
       FIG. 7  is a diagram illustrating a data configuration example of the correction factor table  32 . As illustrated in  FIG. 7 , the correction factor table  32  includes frequency, amplitude ratio, and phase difference fields. 
     The frequency field stores the frequency corresponding to the subcarrier. The respective fields of amplitude ratio and phase difference store the amplitude ratio and the phase difference of the test signal in the frequency of the frequency field, which are obtained when the DU ratio falls to or below the threshold value. 
     For example, it is understood from the correction factor table  32  that, in the example of  FIG. 7 , A l  and θ l  respectively represent the amplitude ratio and the phase difference of the test signal in a frequency f l  corresponding to Subcarrier No. l, which are obtained when the DU ratio falls to or below the threshold value. 
     The compensation for the IQ imbalance will be described in detail. 
       FIG. 8  is a block diagram of the transmission signal generation unit  31  illustrated in  FIG. 2 . As illustrated in  FIG. 8 , the transmission signal generation unit  31  includes a serial-parallel conversion unit  71 , subcarrier modulation units  72   a  to  72   n,  a correction factor computing unit  73 , an IFFT  74 , and a parallel-serial conversion unit  75 .  FIG. 8  also illustrates the correction factor table  32 . 
     The serial-parallel conversion unit  71  receives an input of serial transmission data. The serial-parallel conversion unit  71  converts the input serial transmission data into parallel data, and places the data on the frequency axis (subcarriers f 0 , f 1 , . . . , and f N-1 ). 
     The subcarrier modulation units  72   a  to  72   n  map the transmission data placed by the serial-parallel conversion unit  71  onto signal points of, for example, the QPSK or 16QAM constellation. 
     The correction factor computing unit  73  applies the correction factor stored in the correction factor table  32  to the signal subjected to the subcarrier modulation (primary modulation). For example, the correction factor computing unit  73  applies an amplitude ratio A 0  and a phase difference θ 0  of the correction factor table  32  in  FIG. 7  to the signal output from the subcarrier modulation unit  72   a.  Further, the correction factor computing unit  73  applies an amplitude ratio A 1  and a phase difference θ 1  of the correction factor table  32  in  FIG. 7  to the signal output from the subcarrier modulation unit  72   b.    
     The IFFT  74  performs an inverse Fourier transform on the signal applied with the correction factor by the correction factor computing unit  73 . That is, the IFFT  74  converts the signal in the frequency domain allocated to the subcarrier into a signal sequence in the time domain. 
     The parallel-serial conversion unit  75  converts the signal sequence in the time domain output in parallel from the IFFT  74  into serial data, and outputs the serial data. In this process, the insertion of guard intervals is performed. 
     As described above, the transmission data is placed on the frequency axis by the serial-parallel conversion unit  71 , and is subjected to the primary modulation by the subcarrier modulation units  72   a  to  72   n  in accordance with the QPSK or 16QAM system, for example. The transmission data subjected to the primary modulation is represented by the following equation (11). 
       d l =R l e jφl    (11)
 
     Herein, l represents the subcarrier number (l=0, 1, . . . , or N-1), and d l  represents the transmission data subjected to the primary modulation. Further, R l  and φ l  represent the amplitude and the phase, respectively. The transmission data d l  is mapped on a complex plane, and is represented as d l =1+j in the QPSK system, for example. 
     Subjected to IFFT, the above transmission data is converted into a transmission signal on the time axis. The transmission signal is represented by IDFT (Inverse Discrete Fourier Transform), as in the following equation (12). 
     
       
         
           
             
               
                 
                   
                     S 
                      
                     
                       ( 
                       k 
                       ) 
                     
                   
                   = 
                   
                     
                       ∑ 
                       
                         l 
                         = 
                         0 
                       
                       
                         N 
                         - 
                         1 
                       
                     
                      
                     
                       
                         d 
                         l 
                       
                        
                       
                          
                         
                           j 
                            
                           
                               
                           
                            
                           2 
                            
                           π 
                            
                           
                               
                           
                            
                           l 
                            
                           
                             k 
                             N 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   12 
                   ) 
                 
               
             
           
         
       
     
     Herein, N represents the number of points in the IFFT, and k represents the sampling point of the transmission signal on the time axis (k=0, 1, . . . , or N-1). 
     Herein, the signal subjected to the primary modulation and represented by the equation (11) is multiplied by the amplitude of the correction factor table  32  by the correction factor computing unit  73 , and is added with the phase. Therefore, the signal output from the correction factor computing unit  73  is represented as in the equation (13). 
         {tilde over (d)}=A   l   R   l   e   j(φ     l     +θ     l     )    (13)
 
     With this corrected signal subjected to an inverse Fourier transform by the IFFT  74 , it is possible to obtain a transmission signal on the time axis for compensating for the IQ imbalance, which is represented by the following equation (14). 
     
       
         
           
             
               
                 
                   
                     
                       S 
                       ~ 
                     
                      
                     
                       ( 
                       k 
                       ) 
                     
                   
                   = 
                   
                     
                       ∑ 
                       
                         l 
                         = 
                         0 
                       
                       
                         N 
                         - 
                         1 
                       
                     
                      
                     
                       
                         
                           d 
                           ~ 
                         
                         l 
                       
                        
                       
                          
                         
                           j 
                            
                           
                               
                           
                            
                           2 
                            
                           π 
                            
                           
                               
                           
                            
                           l 
                            
                           
                             k 
                             N 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   14 
                   ) 
                 
               
             
           
         
       
     
     The radio communication device is thus configured to output the test signals to the LPFs  13   a  and  13   b,  return the test signals to the reception unit via the RF unit of the IQ modulation unit  14  and the PA  15 , and calculate the correction factor. Accordingly, it is possible to compensate for the IQ imbalance of the LPFs  13   a  and  13   b  and the RF unit. 
     Further, the test radio signal is down-converted into the IF, and is subjected to quadrature demodulation by the DDC  36 . Accordingly, it is possible to calculate an appropriate correction factor without requiring the IQ demodulation unit  21  to achieve highly accurate orthogonality. 
     Further, with the correction factor calculated upon power-on of the device or periodically, it is possible to handle a change in IQ imbalance caused by a change in temperature. 
     Further, the test signals are returned within the radio communication device. Therefore, there is no influence of image reception due to space propagation, and the LPFs  22   a  and  22   b  do not require a channel selection filter or the like. 
     In the example of  FIG. 2 , the correction factor table  32 , the test signal generation unit  33 , the DDC  36 , the FFT  37 , and the correction factor calculation unit  38  are included in the baseband signal processing unit  11 . However, these components may be provided outside the baseband signal processing unit  11 . 
     Further, the output of the LNA  19  is provided with the switch  20 . However, the input of the LNA  19  may be provided with the switch  20 . 
     Subsequently, a third embodiment will be described. In the second embodiment, the IF-shifted signal for down-converting the test radio signal into the IF is generated through the frequency shift of the frequency of the local signal by a frequency shifter. In the third embodiment, the IF-shifted signal is generated by an independent oscillator. 
       FIG. 9  is a block diagram of a radio communication device according to the third embodiment. In  FIG. 9 , the same components as those of  FIG. 2  are denoted by the same reference numerals, and description thereof will be omitted. 
     In the radio communication device of  FIG. 9 , as compared with the radio communication device of  FIG. 2 , the frequency shifter  25  and the switch  26  are omitted, and an IF oscillator  81  and a switch  82  are provided. The IF oscillator  81  outputs an IF-shifted signal for down-converting, into the IF, the frequency of the test radio signal input to the IQ demodulation unit  21  via the transmission unit, the ATT  18 , and the switch  20 . The frequency of the IF-shifted signal is represented as ω LO . 
     When the test signals are output from the baseband signal processing unit  11 , the switch  82  performs switching such that the IF-shifted signal output from the IF oscillator  81  is output to the IQ demodulation unit  21 . When the transmission signals to be transmitted to the other communication party are output from the baseband signal processing unit  11 , the switch  82  performs switching such that the local signal of the local oscillator  24  is output to the IQ demodulation unit  21 . 
     With the IF-shifted signal thus output by the IF oscillator  81 , the circuit configuration can be simplified. 
     Subsequently, a fourth embodiment will be described. In the second and third embodiments, a switch for looping back a test pattern is provided to the output of the PA. In the fourth embodiment, the input of the PA is provided with a switch for looping back the test radio signal. 
       FIG. 10  is a block diagram of a radio communication device according to the fourth embodiment. In  FIG. 10 , the same components as those of  FIG. 9  are denoted by the same reference numerals, and description thereof will be omitted. 
     In the radio communication device of  FIG. 10 , as compared with the radio communication device of  FIG. 9 , the respective positions of a switch  91  and a PA  92  are reversed. That is, the switch  91  is provided between the IQ modulation unit  14  and the PA  92 . 
     With the switch  91  thus provided at a previous stage of the PA  92 , it is possible to compensate for the loss of the radio transmission signal in the switch  91  with the gain of the PA  92 . 
     The above-described compensation techniques, respectively, can reduce, if not substantially eliminate, IQ (In-phase/Quadrature) imbalance. 
     All examples and conditional language recited herein are intended for pedagogical purposes to aid the reader in understanding the principles of the invention and the concepts contributed by the inventor to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a showing of the superiority and inferiority of the invention. Although the embodiment(s) of the invention(s) has(have) been described in detail, it should be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention.