Patent Publication Number: US-11394378-B2

Title: Power switch with an integrated temperature and current sense circuit

Description:
TECHNICAL FIELD 
     The present invention relates generally to a system and method for a power switch with an integrated temperature and current sense circuit. 
     BACKGROUND 
     A power semiconductor device is a semiconductor device that may be used as a switch or rectifier for power electronics. Power semiconductor devices, which may be referred to as power devices, are often formed as integrated circuits (“ICs”) to produce power ICs. The applications of power devices are numerous and advances in technology have further increased the number of possible applications, especially in the field of power ICs. 
     Power devices are most commonly implemented as power switches in order to operate in either a conduction mode (ON) or a non-conduction mode (OFF). Often power devices are used to block a large voltage from being supplied to a load or to supply the large voltage across the load. 
     Some common power devices are the power diode, thyristor, power metal-oxide-semiconductor field effect transistor (“MOSFET”), and insulated gate bipolar transistor (“IGBT”). 
     Due to the increased current or voltage generally associated with power devices, a power device is often structurally designed in order to accommodate the higher current density, higher power dissipation, or higher breakdown voltage. For example, power devices are often built using a vertical structure and have a current rating proportional to the device&#39;s area and a voltage blocking capability related to the height or thickness of the device in the substrate. With vertical power devices, as compared to lateral non-power devices, one of the device terminals is located on the bottom of the semiconductor die. 
     Power devices sometimes include current and temperature sensing mechanisms to monitor overcurrent or over temperature operating conditions. The output of such sensing mechanisms can be passed to other control and protection circuits for controlling the operation of one or more power devices. Such control and protection circuits operate to disable power devices when an overcurrent or over temperature operating condition is detected. 
     SUMMARY 
     In accordance with an embodiment, an integrated circuit comprises a power switch comprising a current path and a current sense node, and a temperature sense circuit internally coupled between the current path and the current sense node. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1A  is a schematic of a power switch (in this example drawn as IGBT with antiparallel diode) with a current sense and a temperature sense circuit according to an embodiment; 
         FIG. 1B  is a plan view of a power switch layout for the power switch of  FIG. 1A ; 
         FIG. 1C  is a schematic of a power switch and a temperature sense circuit according to an embodiment; 
         FIG. 1D  is a schematic of a power switch with higher inductance between auxiliary and load emitter showing an Electric Over Stress (“EOS”) event in transient load switching events; 
         FIG. 1E  is a schematic of a power switch (in this example drawn using MOSFET transistors), but without the previously described accompanying diode and sense circuits according to an embodiment; 
         FIG. 2A  is a schematic of a power switch and a temperature sense circuit according to an embodiment; 
         FIGS. 2B and 2C  are plan views of power switch layouts for the power switch of  FIG. 2A ; 
         FIGS. 3 and 4  are schematic diagrams of the power switch shown in  FIG. 2A  including additional circuitry resident on a printed circuit board; 
         FIG. 5  is a schematic diagram of the power switch shown in  FIG. 3  further illustrating transient switching operating conditions including parasitic inductances in the load emitter path; 
         FIGS. 6A, 6B, and 6C  are plan views of power switch layouts for the power switches shown in  FIGS. 1A and 2A  further illustrating top-side emitter bonding pad placement; 
         FIG. 7  is a schematic diagram of the power switch shown in  FIG. 2A  further including additional circuitry used for a test mode of operation; 
         FIG. 8  is a flow chart for self-check and Pulse Width Modulation (“PWM”) modes of operation for the power switch embodiments of  FIGS. 2A and 7 ; and 
         FIG. 9  is a timing diagram showing the relative times at which a current sensing mode and a temperature sensing mode are performed relative to a PWM input waveform. 
     
    
    
     DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
       FIG. 1A  shows a schematic of an integrated circuit  100 A according to an embodiment. The integrated circuit  100 A includes an IGBT  102  having a collector coupled to a collector node C and an optional auxiliary collector node Ca, a gate coupled to a gate node G, and an emitter coupled to an emitter node E and an optional auxiliary emitter node Ea. The auxiliary collector and emitter nodes Ca, Ea may be dictated, for example, by the form factor of a package used for the integrated circuit  100 A. The auxiliary collector node Ca and auxiliary emitter node Ea are typically used in applications where the corresponding collector and emitter power switch voltage potentials (at collector node C and emitter node E) have to be measured precisely and/or the power switch has to be switched quickly, although the power chip (integrated circuit) is implemented in an environment with non-negligible stray inductances in the load current paths (e.g. several nH stray inductance). Integrated circuit  100 A also provides a current mirroring function, wherein a sense emitter provides a small portion of an emitter current of the IGBT  102  at a sense emitter node S. Typically, the ratio of a sense emitter current (provided by a small number of sense emitter cells) to the emitter current (provided by a relatively large number of emitter cells) can be about 1/1000, but the ratio can vary for a particular application. An anti-parallel freewheeling power diode  104  is also included in integrated circuit  100 A and is coupled between the collector node C and the emitter node E. As also shown in  FIG. 1A , the integrated circuit  100 A further comprises an on-chip temperature sense circuit Ts comprising a diode stack  110  and an anti-parallel diode  112  coupled between nodes Ta and Tk. In  FIG. 1A , three diodes are shown in diode stack no, but any number can be used. The use of the several diodes in series is done to assure greater accuracy in measuring the voltage across the diodes, which in turn assures greater accuracy in determining the temperature of the integrated circuit  100 A. The voltage of the diode stack  110  can be converted to temperature according to equations that are known in the art. The anti-parallel diode  112  is used to clamp a voltage at emitter node E of IGBT  102 , as will be described in further detail below. 
     The detailed operation of temperature and current sense circuits of integrated circuit  100 A is described in further detail below. The electrical connections of nodes Ta and Tk of the temperature sense circuit Ts can be configured in various ways according to embodiments. 
     An example IGBT integrated circuit layout  100 B of the IGBT integrated circuit  100 A is shown in  FIG. 1B . The gate node G, the sense emitter node S, the auxiliary emitter node Ea, and the temperature sense circuit nodes Tk and Ta are shown as bonding pads arranged in a line on a top surface of the IGBT integrated circuit layout  100 B. The emitter node E is shown as being configured as three power stripes on the top surface of the IGBT integrated circuit layout  100 B, and the temperature sense circuit Ts occupies a small portion of a central one of the three power stripes of the emitter node E shown in  FIG. 1B . Since IGBT  102  is a vertical device, the collector node C occupies the entire bottom surface of the layout  100 B, with only a small edge portion being visible from the top surface of IGBT integrated circuit layout  100 B. IGBT integrated circuit layout  100 B also corresponds to a physical embodiment of integrated circuit  100 A, wherein integrated circuit  100 A comprises a semiconductor body having a top surface and a bottom surface, wherein the top surface of IGBT integrated circuit layout  100 B corresponds to the actual top surface of integrated circuit  100 A and the bottom surface of IGBT integrated circuit layout  100 B corresponds to the actual bottom surface of integrated circuit  100 A. The integrated circuit and layout embodiment described above, as well as other integrated circuit and layout embodiments described herein therefore correspond to an equivalent physical embodiment of an integrated circuit having a semiconductor body with a top surface and a bottom surface. 
     The embodiment of integrated circuit  100 A shown in  FIG. 1A  has maximum flexibility in the interconnection of the circuit nodes. Integrated circuit layout  100 B has five signal pads as previously described, which leads to a high connection effort and a low package utilization. ElectroStatic Discharge (“ESD”) robustness is low as the temperature sense diodes  110  and  112  have a floating potential as previously described. ESD robustness of the sense emitter node S is low, especially when a low number of current sense IGBT cells are implemented. However, a high number of current sense IGBT cells to improve ESD robustness leads to a lower active area utilization. The current mirror IGBT cells ideally have low external inductance since they are typically connected to a sensing resistor or equivalent circuit. If the sense emitter node S has high stray inductance, IGBT destruction from corresponding high voltages can result during transient switching operations. Such an event can be generally described as an Electric Over Stress (EOS) event. 
     Due to the ESD weakness of the temperature sense diodes  110  (note again that both nodes Ta and Tk are floating in the integrated circuit  100 A of  FIG. 1A ), one branch of the temperature sense diodes  110  can be coupled to the emitter potential of the IGBT  102 . IGBT integrated circuit  100 C is thus shown as having coupled Tk and Ea nodes in an embodiment. The ESD robustness of the embodiment of IGBT integrated circuit  100 C shown in  FIG. 1C  is thus improved compared to the IGBT integrated circuit  100 A shown in  FIG. 1A . The Tk node of the temperature sense circuit Ts does not have a floating potential in integrated circuit  100 C. 
     The stray inductance problem described above is shown in further detail in integrated circuit  100 D of  FIG. 1D . The IGBT  102  and anti-parallel diode  104  are depicted in  FIG. 1D  as well as an external load resistor Rl coupled between the emitter and auxiliary emitter nodes E and Ea, as well as an external sense resistor Rs coupled between the sense emitter and emitter nodes S and E. The temperature sense circuit is not shown in  FIG. 1D . A problem in an application of integrated circuit  100 D is that the current mirror cells (current sense IGBT cells) can be damaged by high voltage transients. In  FIG. 1D , the sense emitter node S is connected to the load path with higher stray inductance through sense resistor Rs. During switching, high voltage transients and can lead to the gate voltage exceeding the maximum allowed gate-to-source voltage of the current sense IGBT cells (Vgs), which is generally described herein as an Electric Over Stress (EOS) event. 
     While an IGBT  102  is shown in the integrated circuit  100 A of  FIG. 1A , a power MOSFET transistor or transistors can be substituted as well, including a current path between a drain node and one of at least two source nodes, wherein the current flow through the current path is controlled by a gate node. One of the at least two source nodes can be used to provide the current mirroring function in a similar manner as described above to provide a small portion of current flow, while the other one of the at least two source nodes is configured to provide a major portion of the current flow as a load current to a load in an on-state of the power MOSFET transistor. Referring now to  FIG. 1E , an example MOSFET power switch  140  is shown including a first MOSFET transistor  120  having a device area of “N” area units and a second MOSFET transistor  122  having a device area of “1” area unit. The drains of MOSFET transistors  120  and  122  are coupled together at a common drain node  142 , and the gates of MOSFET transistors  120  and  122  are coupled together at a common gate node  144 . Note that a first source node  146  provides the majority of the current flow in the power switch  140  as, for example, a load current. A second source node  148  provides a small portion of the current flow in the power switch  140  as, for example, a sense current. The sense current and the load current are ratioed according to the respective device areas. The sense and diode circuits are not depicted in  FIG. 1E , but, if used, would be substantially the same as previously described with reference to  FIG. 1A . Similar substitutions of IGBT and MOSFET power switches can be made for embodiments described below, wherein only an IGBT implementation may be shown. 
     Further power switch embodiments are discussed below that address the above performance and utilization issues. 
     An embodiment of a power switch realized as an integrated circuit  200 A is shown in  FIG. 2A . The IGBT  102  and the freewheeling diode  104 , as well as the corresponding nodes are substantially the same as previously described. However, the on-chip temperature sense diodes  110  and  112  of temperature sense circuit Ts are internally connected between the sense emitter node S and the auxiliary emitter node Ea or emitter node E. The sense emitter cells of IGBT  102  and temperature sense diodes  110  and  112  are designed in such way that the temperature sense diodes can carry all of the current of the sense emitter cells in case the sense emitter node S is externally unconnected. The ability to carry all of the current at sense emitter node S is achieved either by a strong enough (large enough area) of temperature sense diodes  110  and  112  or by a low number (to reduce current flow) of sense emitter cells in IGBT  102 , or both. The benefits and operation of the integrated circuit  200 A shown in  FIG. 2A  and related embodiments are discussed in further detail below. 
     Embodiment layouts  200 B and  200 C of integrated circuit  200 A are shown in  FIGS. 2B and 2C . It is not mandatory to have a physically separate Ea pad as is shown in layout  200 B of  FIG. 2B . Layout  200 B includes bonding pads only for the gate node G and sense emitter node S. The bonding pad of the auxiliary emitter node Ea is merged into the center one of the three emitter power stripes E as shown. As in the embodiment of  FIG. 1B , the temperature sensing circuit Ts occupies a portion of one of the emitter power stripes E. The collector node C is on the bottom side of the integrated circuit, and a small edge portion of the collector node C is shown in  FIG. 2B . 
       FIG. 2C  shows an alternative layout  200 C for integrated circuit  200 A that uses a dedicated bonding pad at auxiliary emitter node Ea. The bonding pads for gate node G and sense emitter node S, as well as the emitter power stripes of emitter node E, and the collector node C are substantially the same as in layout  200 B shown in  FIG. 2B . 
     The temperature sense circuit Ts can be placed in the center of the layout  200 B, as is shown in  FIG. 2B  for precise integrated circuit temperature measurements, or on the side of the emitter power stripes of emitter node E for minimizing the loss of active area of the IGBT load area as shown in the layout  200 C of  FIG. 2C . While two locations for temperature sense circuit Ts are shown in  FIGS. 2B and 2C , other integrated circuit locations can be used as well. It is also suitable to place the temperature sense diodes distributed over the chip area in order to get an averaged signal of, e.g., the chip center and the chip corner. 
     The layouts  200 B and  200 C shown in  FIGS. 2B and 2C  use a low number of signal pads (one pad for gate node G, one pad for sense emitter node S, and an optional pad for auxiliary emitter node Ea), which leads to minimum package effort for accessing the bond pads and to high active area utilization. The layouts  200 B and  200 C shown in  FIGS. 2B and 2C  have the highest ESD robustness as no areas of the integrated circuit are floating. Also the small current mirror function provided by IGBT  102  is protected via the temperature sense diodes  110  and  112 . The layouts  200 B and  200 C also have high EOS robustness in case high stray inductance loops are externally connected to a sense resistor external component (Rs) described in further detail below. 
       FIG. 3  shows a power switch realized as an integrated circuit  200 D including additional external circuitry needed to complete the temperature sensing and current sensing functionality. Integrated circuit  200 D is coupled to an external sense resistor Rs and an external transistor  106  (having a smaller size than a size of a power transistor) resident on a printed circuit board (PCB) that are not integrated into integrated circuit  200 D, in an embodiment. Sense resistor Rs and external transistor  106  are coupled between sense emitter node S and auxiliary emitter node Ea, in an embodiment. A gate node G 1  of IGBT  102  is coupled to a gate node G 2  of transistor  106 , in an embodiment. Transistor  106  is shown as a MOSFET in an embodiment, but other types of transistors can be used. 
     In operation, when the coupled gate nodes G 1  and G 2  are both switched high, the current of IGBT  102  can be sensed (current sensing method) by measuring the voltage across sense resistor Rs at the sense emitter node S and auxiliary emitter node Ea, dividing by the resistance of resistor Rs, and multiplying by the ratio of the emitter size to the sense emitter size. When the coupled gate nodes G 1  and G 2  are both switched low, the temperature of IGBT  102  can be sensed (temperature sensing method) by injecting a current into the sense emitter node S and auxiliary emitter node Ea, measuring the developed voltage at the sense emitter node S and the auxiliary emitter node Ea, dividing by the number of diodes used, and interpolating the voltage to a diode temperature using diode current equations known to those in the aft. 
     As described above, the transistor  106  can be controlled with the same gate voltage as the load IGBT  102 . Then automatically, when IGBT  102  is turned on the integrated circuit  200 D is in the current sensing mode. An advantage of integrated circuit  200 D is that if stray inductance of the sense emitter node S results in high voltage overshoots, these overshoots will be clamped by the temperature sense diodes  110  and  112  thus protecting the IGBT  102  from EOS events and destruction. As described above, when the IGBT  102  is turned off the sensing resistor Rs is disconnected and the temperature sense diodes  110  and  112  can be used for temperature sensing. 
     A general advantage of integrated circuits  200 A and  200 D shown in  FIG. 2A  and  FIG. 3  is that the IGBT  102  is connected/tested with only external pads associated with the gate node G, collector node C, and emitter node E, even though all other pads (pins) are externally floating. However, even though the pads of sense emitter node S, auxiliary collector node Ca, and auxiliary emitter node Ea may be floating, no critical potentials will occur due to these open pads because of the internal circuitry and corresponding electrical connections as previously described. In contrast, in integrated circuit  100 A of  FIG. 1A , at least some of these pads must be connected externally in order to prevent floating nodes and to overcome the ESD as well as EOS problems previously described. 
       FIG. 4  shows an integrated circuit  200 E and additional external circuitry configured for the temperature sensing mode. A current source  108  injects current to the temperature sensing diodes  110  and measures via the pn junction drop the temperature of the IGBT  102 . In the temperature sensing mode, the gate nodes are OFF. The description of the integrated circuit  200 E is similar to that of the description of the integrated circuit  200 D previously described with respect to  FIG. 3 , except for the addition of the current source  108  that is also an external component and not integrated together with integrated circuit  200 E. 
     Additionally, related to the temperature sensing mode, a diagnosis of sense resistor Rs and the current source  108  can also be performed, if desired. The diagnosis is performed in a no load operating mode or in a freewheeling operating mode where the antiparallel diode  104  conducts current and no current flows through the IGBT  102 . The IGBT  102  can be turned-on in the diagnosis mode as long as there is no load current. The high gate value also turns on transistor  106  and also connects the sensing resistor Rs between the sense emitter node S and the auxiliary emitter node Ea. Due to the R=U/I ohmic law it is now possible with a known sense resistor value (R) and a measured voltage drop (U) to check the value of the small signal current source (I). Or, alternatively, when the current source has a known status the sense resistor value Rs can be checked. A diagnosis mode is thus possible during application time for current source  108  and/or sense resistor Rs, which can show also degradation, drift failures over lifetime and can lead to wrong signals. 
     In operation, the current source diagnosis proceeds according to the following steps: 
     1. During operation, wait for a no load, no switching operational phase (in the alternative, a freewheeling phase during switching). In this operation phase, the current emitter does not inject current into temperature sense circuit Ts or sensing resistor Rs. 
     2. Enable current source  108  and the sensing resistor Rs by turning on the auxiliary gate G 2  of transistor  106 . 
     3. Measure the voltage drop across sense emitter node S and auxiliary emitter node Ea. 
     4.1. With a known value of sensing resistor Rs (including the parasitic MOSFET resistance of transistor  106 ), the current source injects a current I=Voltage_S_Ea/Rs, wherein the term Voltage_S_Ea represents the voltage across the sense emitter node S and auxiliary emitter node Ea. This measured current value can be compared with the set point of the current source. 
     4.2. With a known current source current, the value of sensing resistor Rs (including the parasitic MOSFET resistance of transistor  106 ) can be calculated as Rs=Voltage_S_Ea/Isource. This resistance value can be compared with the implemented resistance value and thus the sense resistor Rs can be periodically checked for drifts over the lifetime of the integrated circuit  200 E and/or degradation or damage of the sensing resistor Rs. 
     The inverter system (not shown) utilizing the integrated circuit and supporting circuitry of described embodiments will be more robust against ESD events and also for electric overstress (EOS). Integrated circuit  100 D of  FIG. 1D  showed an embodiment where the current sense is connected to a relative high stray inductance in the load path. The stray high inductance can lead to IGBT failures as was previously described. 
     The integrated circuit  200 F and supporting circuitry shown in  FIG. 5  protects the integrated circuit from Vgs overvoltages in the case of high inductance connections. While circuit  200 F has been previously described with respect to circuit  200 D of  FIG. 3 , it is reproduced here with additional directional arrows to further illustrate the EOS robustness improvement for high inductance connection applications. 
     With the circuit  200 F shown in  FIG. 5 , the internal connected temperature sense diodes  110  and  112  in the temperature sensing circuit Ts connects the sense emitter node S to the auxiliary emitter node Ea. In the case of transient higher voltages on the emitter node E, the temperature sense diodes  110  and  112  will clamp or limit the overvoltage and thus protect the IGBT  102  from an electric overstress event (“EOS”). When a customer has open sensor pads in an application, the same principle will protect the IGBT from overvoltage and makes the integrated circuit more robust. Thus,  FIG. 5  depicts a protected value of Vgs between the gate node G 1  and sense emitter node S due the voltage clamping action of diodes  110  in a first direction, and the voltage clamping action of diode  112  is a second direction. 
       FIGS. 6A, 6B, and 6C  further illustrate integrated circuit area utilization according to embodiments. Particularly, the utilization of the top-side emitter bonding wires are shown. In  FIG. 6A , a layout  100 E of an IGBT circuit including a temperature sensing circuit Ts is shown wherein separate gate node G, sense emitter node S, auxiliary emitter node Ea, and temperature sensing circuit nodes Tk and Ta are brought out to individual bonding pads. In this embodiment, there are eighteen bonding locations available coupled to six emitter bonding wires  602 . In  FIG. 6B , a layout  200 G of an IGBT circuit including a temperature sensing circuit Ts is shown wherein only a separate gate node G, a sense emitter node S, and an auxiliary emitter node Ea are brought out to individual bonding pads. In this embodiment, note that there are an additional three bonding locations available with an additional two emitter bonding wires. In total, there are eighteen bonding locations (in layout  100 E) with eight emitter bonding wires  604 . In  FIG. 6C , a layout  200 H of an IGBT circuit including a temperature sensing circuit Ts is shown wherein only a separate gate G and a sense emitter node S are brought out to individual bonding pads. In this embodiment, note that there are an additional six bonding locations available in additional to the eighteen bonding locations (in layout  100 E) for coupling to bonding wires  606 . Thus,  FIGS. 6B and 6C  illustrate in additional detail that there is greater integrated circuit area available for power distribution when fewer bonding pads are used according to the embodiments described herein. 
       FIG. 7  shows an integrated circuit  200 I that is suitable for use in a self-test mode that is further described in greater detail below. Circuit  200 I is the same as was previously described with integrated circuit  200 E of  FIG. 4 , except that the gate G 1  of IGBT  102  is separately and independently operable from the gate G 2  of transistor  106 . 
     When the gate G 1  is off and gate G 2  is on, a self-test of the temperature sensing circuit Ts can be active using current source  108 . Due to a known defined current for current source  108 , it is possible to check the sense resistor value Rs as well as the function of the additional transistor  106 . In a case where the current source  108  has a malfunction, this can also be detected. The sense resistor Rs and current source  108  can thus be periodically tested as desired during the functional life of the IGBT or power switch circuit. 
     A flow chart  800  shown in  FIG. 8  sets forth a method of operating a circuit including an on-chip temperature sensing circuit according to embodiments including a self-test or self-check mode in addition to a normal PWM operating mode. In the self-check mode, gate G 1  and gate G 2  are controlled individually. The individual control enables turning off IGBT  102  and checking with the current source  108  the current sense resistor Rs connection as well as the function of the current source  108 . In normal PWM operation, gate G 1  and gate G 2  are controlled with same signal. 
     In flow chart  800 , the method of operating the circuit including an on-chip temperature sensing circuit starts at step  802 . A self-check is entered into first at self-check mode step  804 . In self-check mode step  804 , gate G 1  is off, gate G 2  is on, and current source is on with a nominal value of one milli-amp. The nominal value of one milli-amp is given as an example only and another nominal value of current source could be used. At check voltage step  806 , the voltage Voltage_S_Ea between the sense emitter node S and the auxiliary emitter node Ea (or emitter node E if an auxiliary emitter node is not used) is checked. If the voltage is not within an expected voltage range  810 , then an error flag is raised at step  808 . In a production mode, the tested IGBT circuit can be determined to be faulty and reworked or scrapped. In an application mode, the IGBT circuit can be shut down by the surrounding operating system. If the voltage is within an expected voltage range  810 , the method proceeds to a PWM operation mode at step  812 . 
     An example expected voltage range  810 , which is used at check voltage step  806 , is given by the following equations:
 
Voltage_ S _ Ea &lt;CurrentSource*( Rs +RauxTransistor)+Threshold(5%) and
 
Voltage_ S _ Ea &gt;CurrentSource*( Rs +RauxTransistor)−Threshold(5%),
 
     wherein Voltage_S_Ea was previously defined, CurrentSource is the current value of current source  108 , Rs is the resistance value of the sense resistor Rs, RauxTransistor is the series resistance value of transistor  106 , and Threshold(5%) is a threshold voltage value that is five percent (as an example only) of an expected voltage value of Voltage_S_Ea. 
     At step  812 , normal PWM operation is started. In normal PWM operation, gate G 1  and gate G 2  are coupled together to a common gate voltage, either ON or OFF. If the gates G 1  and G 2  are both off, a temperature sensing is performed at step  814 , and, if the gates G 1  and G 2  are both on, then a current sensing is performed at step  816 . 
     In the temperature sensing mode at step  814 , current source  108  has a nominal value of one milli-amp and the voltage Voltage_S_Ea is measured. The measured voltage Voltage_S_Ea is converted to a temperature reading as previously discussed. 
     In the current sensing mode at step  816 , the voltage Voltage_S_Ea is measured. The measured voltage Voltage_S_Ea is compared to an overcurrent threshold value. If the comparison is normal, the PWM mode continues normally. If the comparison indicates an overcurrent condition, then the IGBT circuit can be turned off or a warning signal can be generated, or other actions can be taken to minimize power dissipation and thus damage to the IGBT. 
     The interaction of the current sensing modes and the temperature sensing modes during normal PWM operation is further explained with reference to the timing diagram  900  of  FIG. 9 . Temperature sensing and current sensing are synchronous with the control pattern of the PWM signal. The temperature signal is sampled during the PWM off phases, and ideally center aligned in the PWM signal in middle of the off phase. 
     At counter zero, the power switch is in the middle of a freewheeling phase and ideally the temperature signal is sampled during this phase. During the normal PWM mode, the current sensing circuit is activated with a continuous comparing of the measured current signal to a fault threshold during entire turn-on mode. In the case of a current signal being recorded, it will be done at the maximum counter value. Temperature sensing is done at the minimum counter value in the center of freewheeling mode. The temperature signal is typically sampled with Analog-to-Digital (“ADC”) circuits (not shown). The sampling is preferably triggered a short time after the gate is turned-off after switching transients and noise disappears or alternatively in the center aligned PWM at middle of the freewheeling phase (i.e. at PWM counter zero). 
     Thus, the timing diagram  900  of  FIG. 9  shows a PWM counter signal alternating between a zero count value and a maximum count value. The combined gates G 1  and G 2  are ON if the counter value  902  is above a PWM comparison threshold  904 , otherwise the gates G 1  and G 2  are OFF as is shown by gate timing signal  906 . When the gates are ON between times t 1  and t 2 , and t 3  and t 4 , the current sensing is enabled as is shown by current sense signal  908 . When the gates are OFF between times t 2  and t 3 , the temperature sensing is enabled as is shown by temperature sense signal  910 . Triggering of the temperature sensing occurs either a short time after t 2  or at the center of the OFF mode at the minimum count value (at a time of (t 3 −t 2 )/2)). 
     In summary, a power semiconductor (which can be a MOSFET, IGBT, Gate Turn-Off thyristor (“GTO”), or other type of power device) with an on-chip current and temperature sensor internally coupled (“internal” and “internally” being defined as coupling inside of the integrated circuit and not with a device or devices and corresponding connections that are external to the integrated circuit) between the sense emitter node S and the auxiliary emitter node Ea or emitter node E has been described. The on-chip current sensor can be implemented as providing a current mirror function with a split emitter arrangement as described or with separate pn-diode structures if desired. The temperature sensor is implemented with anti-parallel diodes that are connected between the current sensor (sense emitter) and the load pad (emitter). Due to this internal connection the ESD and Electrostatic OverStress (“EOS”) robustness is improved. Furthermore, the number of signal pads is minimized leading to higher utilization of active integrated circuit area. 
     In an embodiment, an integrated circuit comprises a power switch comprising a current path and a current sense node; and a temperature sense circuit internally coupled between the current path and the current sense node of the power switch. The current path comprises a drain node and a first source node, and the current sense node comprises a second source node in an embodiment. The current path comprises a collector node and an emitter node, and the current sense node comprises a sense emitter node in an embodiment. The temperature sense circuit comprises a plurality of serial-connected diodes and an additional diode coupled in anti-parallel with the plurality of serial-connected diodes in an embodiment. The circuit further comprises a current sense resistor and an additional switch coupled between the current sense node and the current path, wherein the power switch further comprises a first control node and the additional switch comprises a second control node coupled to the first control node of the power switch in an embodiment. The temperature sense circuit is configured to be operated in a first mode of operation, wherein the first mode of operation comprises a free-wheeling mode in an embodiment. The circuit is configured to be operated as a current sensing circuit in a second mode of operation, wherein the second mode of operation comprises an on-state of the power switch in an embodiment. 
     In another embodiment, an integrated circuit comprises a power switch comprising a first current node, a second current node, and a third current node; and a temperature sense circuit coupled between the second current node and the third current node of the power switch, wherein the first current node comprises a power connection on a bottom surface of the integrated circuit, the second current node comprises a plurality of power connection stripes on a top surface of the integrated circuit, and the third current node comprises a first bonding pad on the top surface of the integrated circuit. The power switch further comprises a control node comprising a second bonding pad on the top surface of the integrated circuit in an embodiment. The temperature sense circuit is configured to be operated in a free-wheeling mode of operation in an embodiment. The circuit is configured to be operated as a current sensing circuit in an on-state of the power switch in an embodiment. The first current node comprises a collector node, the second current node comprises an emitter node, and the third current node comprises a sense emitter node in an embodiment. The first current node comprises a drain node, the second current node comprises a first source node, and the third current node comprises a second source node. The temperature sense circuit comprises a plurality of serial-connected diodes and an additional diode coupled in anti-parallel with the plurality of serial-connected diodes in an embodiment. 
     In another embodiment, a method comprises configuring a power switch to include a current path and a current sense node; and internal to the power switch, coupling a temperature sense circuit between the current path and the current sense node of the power switch. The method includes determining a temperature of the temperature sense circuit in a first mode of operation, and determining a current flowing in the current sense node in a second mode of operation, according to embodiments. 
     While this invention has been described with reference to illustrative embodiments, this description is not intended to be construed in a limiting sense. Various modifications and combinations of the illustrative embodiments, as well as other embodiments of the invention, will be apparent to persons skilled in the art upon reference to the description. It is therefore intended that the appended claims encompass any such modifications or embodiments.