Patent Publication Number: US-7719595-B1

Title: Preview mode low resolution output system and method

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
   This application is a continuation of U.S. patent application Ser. No. 10/742,170 filed Dec. 19, 2003, now U.S. Pat. No. 7,304,679 which is a continuation application of U.S. patent application Ser. No. 09/282,524 filed on Mar. 31, 1999 entitled “Preview Mode Low Resolution Output System and Method” having inventors Sandra M. Johnson, Douglas R. Holberg, and Nadi R. Itani (Issued as U.S. Pat. No. 6,686,957 and is related to patent application Ser. Nos. 09/283,098; 09/283,112; 09/282,515; 09/283,779; 09/282,523, respectively entitled “Phase Locked Loop Circuits, Systems, and Methods” having inventors Douglas R. Holberg and Sandra Marie Johnson; “CCD Imager Analog Processor Systems and Methods” having inventors Douglas R. Holberg, Sandra Marie Johnson, Nadi Rafik Itani, and Argos R. Cue; “Amplifier System with Reducable Power” having as inventor Nadi Rafik Itani; “Dynamic Range Extender Apparatus, System, and Method for Digital Image Receiver System” having inventors Sandra Marie Johnson and Nadi Rafik Itani, which has issued into U.S. Pat. No. 6,252,536 on Jun. 26, 2001; and “Successive Approximation Apparatus, System, and Method for Dynamic Range Extender” having as inventor Nadi Rafik Itani; each of these applications filed on even date herewith, and each incorporated herein by reference in its entirety. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   This invention relates to analog and digital processors and methods, and more particularly to preview mode low resolution output systems and methods for charge coupled devices (CCDs), CMOS imagers, and cameras. 
   2. Description of the Related Art 
   Camera systems using charge coupled devices (CCDs) and imagers of many kinds are well-known for capturing signals according to many different CCD output formats and pixel configurations. According to one such format, in order to obtain a still image with acceptable resolution and contrast from a CCD, a minimum of 10 bits of resolution is desired. To practically capture a CCD image, the data read-out time from the CCD is very limited. Accordingly, one such front end interface which accepts CCD data for conversion into digital form operations typically up to 16 MHz with a 10-bit analog-to-digital converter. A camera using this front-end can produce a digital still image with up to 8 k×8 k pixels. The feature set available in known CCD camera system is increasing to include more functionality, as well as extended dynamic range. Such extended functionality comes at a price in terms of electronic complexity and power consumption. For example, some current camera systems include a liquid crystal display (LCD) screen to enable viewing of images in a real-time viewfinder. This requires the CCD and associated processing chips to run in a video mode and to remain powered up while the screen is in use. This can dissipate a large amount of power that tends to shorten battery life. In such an operational mode, front end circuitry is operated at a resolution level which is unnecessary for driving the relatively low resolution LCD display, thereby consuming power needlessly. 
   Accordingly, there is a need to enable low power operation of the analog and digital subsystems in CCD camera and imager systems that convert analog data into digital signal forms for user applications. It is desirable to achieve lower power even at a sacrifice in resolution in the front end system. 
   SUMMARY OF THE INVENTION 
   According to one embodiment of the present invention, a processing system for an imager device includes a camera system for producing a desired imager signal which operates in a reduced power or preview mode. Such a system according to the present invention includes a correlated double sample (CDS) circuit for receiving data from a selected imager, a multi-mode (selectably high or low current) variable gain amplifier (VGA), a low power mode analog-to-digital converter (ADC) having a selectable narrow bit-width output and coupled to said CDS circuit. The low power mode enables production of an ADC output signal of selectable higher or lower resolution. The processing system according to the present invention includes a gain adjust block (GAB) coupled to the ADC, a black level adjustment circuit including a predetermined clamp setting, a compander circuit coupled to said GAB for further reducing the output bit-width, a multiplexer permitting selection of output signals of selected bit-width, and a phase-lock-loop (PLL) for controlling a multi-sync timing generator including an analog clock generator (ACG). According to the present invention, the compander bit-width reduction compresses the output so a smaller bit-width signal can retain the same dynamic range as a larger bit-width signal, while the ADC output bit-width reduction sacrifices resolution. According to one embodiment of the present invention, by reducing the resolution requirement of the camera system front end to a selected number of bits during a still camera viewfinder video mode of operation, the power dissipated by the camera system is reduced substantially. In particular according to one embodiment of the present invention, a signal processing system (SPS) on an integrated substrate for a camera has a reduced power preview mode. The camera includes analog front-end (AFE) circuitry with digital outputs selectable for multiple bitwidths and having selectable high and low resolution (preview) output modes, and digital signal processing system (DSPS) circuitry connected to the analog front-end (AFE) circuitry. Further according to the present invention, a signal processing system (SPS) for an imager device includes a camera system for producing an imager signal, a correlated double sample (CDS) circuit for receiving data from an imager, a multi-mode variable gain amplifier (VGA), a low power mode analog-to-digital converter (ADC) having a selectable narrow bit-width output and coupled to said CDS circuit, a digital gain circuit (DGC) coupled to the ADC, and a compander circuit coupled to said DGC for further reducing the output bit-width of the camera system. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a CCD camera system according to the present invention; 
       FIG. 2  is a block diagram of an analog image processing subsystem (AIPS) according to the present invention; 
       FIG. 3  is a diagram of an ideal output waveform of a selected imager, which is processed in accordance with one embodiment of the present invention; 
       FIG. 4A  is a diagram of the transfer function of a VGA circuit according to one embodiment of the present invention; 
       FIG. 4B  is a graph of DOUT as a function of VGA input, with DOUT ranging from zero to 8191, according to one embodiment of the present invention; 
       FIG. 5A  is a block diagram of a correlated double sampling variable gain amplifier (CDS/VGA) for an analog data processing subsystem according to the present invention; 
       FIG. 5B  is a circuit diagram of an amplifier according to an embodiment of the present invention, which is subject to power down performance during a preview mode of operation; 
       FIG. 6  is a timing diagram of the operation of a correlated double sampling variable gain amplifier (CDS/VGA) operating with a two phase clock according to an embodiment of the present invention; 
       FIG. 7A  is a block diagram of an analog-to-digital converter according to one embodiment of the present invention; and 
       FIG. 7B  is a diagram according to the present invention, which shows different levels of resolution output from an ADC, depending upon whether low significant value stages of the ADC are engaged for operation or disengaged. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Referring now to  FIG. 1 , there is shown a block diagram of a camera system  13 , according to the present invention. As shown in  FIG. 1 , camera system  13  according to the present invention includes the following integrated circuit (IC) components, according to one embodiment of the present invention: a CCD array sensor  14 , a vertical driver circuit  15 , first and second signal processing subsystems (SPS)  17  and  18  (i.e., a front-end and a back-end subsystem), a DC-to-DC converter  19 , and a display system such as for example without limitation a liquid crystal display (LCD) panel  20 . The LCD panel  20  is connected to second SPS  18  for receipt of a digital signal input. First SPS  17  is an analog signal processing (ASP) front-end (AFE) system which receives and processes video samples from the CCD array sensor  14  and generates timing clocks and pulses required by the CCD array sensor  14 , and vertical driver circuit  15 . The vertical driver circuit  15  generates high voltage vertical shift register clock signals. The video output of the CCD array sensor  14  is directly connected to the input of the first SPS  17  through an emitter-follower and AC coupling capacitor. DC-to-DC converter  19  receives unregulated 5 volts DC and produces first and second regulated output voltages at 5 and −5 volts. 
   Referring now to  FIG. 2 , there is shown a block diagram of the first signal processing system (SPS)  17  according to the present invention. The Figure particularly shows a block diagram of an analog image processor subsystem (AIPS) referred to generally as front-end in accordance with one embodiment of the present invention. First SPS  17  includes a summation node  43 , a correlated double sampler and variable gain amplifier (CDSVGA) circuit  44  receiving data in the form of an input voltage (VIN) from an image acquisition device (or imager), such as are conventionally known, an analog-to-digital converter (ADC)  46  connected to CDSVGA circuit  44 , a black level adjustment circuit (SLAC)  45  feeding back to the summation node  43 , a gain adjustment circuit  47 , a 13 to 10 bit compressor circuit  48 , and a multiplexer circuit  49  for permitting selection of outputs between the compressor circuit  48  and gain adjustment circuit  47 . Gain adjustment circuit  47  is connected at its input to ADC  46  and at its output to compressor circuit  48 . AIPS  17  additionally includes an analog clock generator circuit  50 , a timing generator circuit  51 , a phase lock loop (PLL) circuit  52 , a reference circuit  53 , a serial interface circuit  54 , and first and second digital-to-analog converters  55  and  56 . Gain adjustment circuit  47  is controlled by CDSVGA circuit. PLL circuit  52  contributes to control of analog clock generator circuit  50 . Timing generator circuit  51  provides timing signals to external circuitry (not shown). Serial interface  54  is connected for communication with black level circuit  45 , analog clock generator  50 , PLL  42 , DAC 1   56 , and DAC 2   57 . 
   Referring now to  FIG. 3 , there is shown a diagram of an ideal output waveform of a selected imager, which is processed in accordance with one embodiment of the present invention. Referring specifically to  FIG. 3 , there is shown a diagram of an ideal output waveform of a selected imager used in connection with the present invention. Correlated double sampling according to the present invention is accomplished by receiving imager output signal which includes reset noise, thermal noise, and 1/f noise, that are generated by the imager. The noise received degrades the S/N ratio and is cancelled by correlated double sampling according to the present invention. Noise received during the active video portion of the CCD signal is assumed to be correlated with the noise originating during the feed-through portion of the signal. This noise is cancelled by subtracting the feed-through level from the video level with correlated double sampling according to the present invention. The active video signal is the difference between feed-through and video levels according to the present invention. The active video signal varies according to light conditions. In order to insure that the full dynamic range of the ADC  46  is utilized even under low light conditions, the imager output is amplified using a variable gain amplifier (VGA). 
   Referring now to  FIG. 4A , there is shown a diagram of the transfer function of a CDS/VGA circuit  44  according to one embodiment of the present invention. In particular,  FIG. 4A  is a graph of the output of CDS/VGA circuit  44  for selected gain settings of 1×-8×, according to one embodiment of the present invention. The Figure expresses the relationship between VGA_OUT and ADC_OUT. Specifically, VGA_OUT=0 maps to code 0 at ADC_OUT and VGA_OUT=full scale maps to code 1023 at ADC_OUT. The ADC_OUTPUT, i.e., the output of the analog-to-digital converter  46 , can range from zero to full-scale (i.e., from code zero to code 1023) while VGA_INPUT values range from zero to about 0.125 at a gain setting of 8×. Alternatively, the output of the analog-to-digital converter  46 , can range from half-scale to full scale (i.e., from code 512 to 1023) when the VGA_INPUT values range from about 0.125 to about 0.25 at a gain setting of 4×. In another case, the output of the analog-to-digital converter  46 , can range from half-scale to full-scale, while the VGA_INPUT ranges from about 0.25 to about 0.5 at a final gain setting of 2×. In another case, the output of the analog-to-digital converter  46 , can range from half-scale to full-scale, while the VGA_INPUT ranges from about 0.5 to about 1.0 at a gain setting of the CDS/VGA  44  of 1×. In operation according to the present invention, the highest possible gain setting is selected for a particular VGA input signal. When a trip point is reached at which the VGA input corresponds to an out-of-range ADC output value, e.g., greater than code 1023, the VGA gain is reduced to a next lower level, which is one half of the immediately prior gain. The trip points lie at regularly spaced intervals from each other, for example at VGA input values which are double the value of the next lower valued trip point. As the VGA input increases in value beyond a particular trip point, the gain of the CDS/VGA  44  is cut in half, resulting in a halved ADC  46  output level. For example, when the ADC output reaches approximately 1023 according to one embodiment, the output level of the ADC  46  abruptly drops to one half of 1023, i.e., approximately to 512, as the gain of the VGA is suddenly cut in half. 
   Referring now to  FIG. 4B , there is shown a graph of the output of the gain adjust block  47  as a function of VGA input, with DOUT ranging from zero to 8191, according to one embodiment of the present invention. The gain adjust block  47  is sued according to the present invention to back out or perform the reverse operation of what is done in the VGA. For example, if a gain of 8 applied in the VGA, the gain adjust block shift the output by 3 bit to the right, thus performing a divide by 8 operation. Thus, whatever the grain is which is applied by the VGA, the gain adjust block applies the inverse of this gain. Accordingly, the output of the gain adjust block contains 13 bits according to one embodiment, and the dynamic range of the 10-bit ADC is increased by 3 bits. To express the DOUT range corresponding to a VGA_INPUT range from zero to a value of about 0.125*full_scale_in, output bits  9 - 0  are employed. To express the DOUT range corresponding to a VGA_INPUT range from 0.2 to about 0.25*full_scale_in, output bits  10 - 1  are employed. To express the DOUT range corresponding to a VGA_INPUT range from 0.25*full_scale_in to about 0.5*full_scale_in, output bits  11 - 2  are employed. To express the DOUT range corresponding to a VGA_INPUT range from 0.5*full_scale_in to about 1.0*full_scale_in, output bits  12 - 3  are employed. As can be seen, the curve of DOUT is smooth, monotonic, and continuous, even at transitions associated with trip points 0.125, 0.25, and 0.5*full_scale_in. The point 1.0*full_scale_in marks the end-of-range for VGA input values, and does not represent a trip point according to this embodiment of the present invention. According to another embodiment of the present invention, in which a 3-bit ADC or an n-bit ADC is used in lieu of an 2-bit ADC, additional thresholds are established within the scope and meaning of the present invention. Such thresholds amount to additional trip points. 
   Referring now to  FIG. 5A , there is shown a block diagram of a correlated double sampling variable gain amplifier (CDS/VGA)  44  for an analog data processing subsystem according to the present invention. Referring particularly to the Figure, there is shown a block diagram of CDS/VGA circuit  44  including first, second, and third CDS/VGA circuit stages respectively  131 ,  132 , and  133 , and a variable capacitor  134  connected to VREF, according to the present invention. First stage  131  includes a first amplifier  136  connected to variable capacitor  134 ; a fixed value capacitor  137  in parallel with first amplifier  136 ; a first switch  138  alternating between open and closed states in accordance with a clock φ 1  in parallel with first amplifier  136 ; and a fixed value input capacitor connected to Vin. According to one embodiment of the present invention, capacitors  137  and  139  have the same capacitance. Second stage  132  of the CDS/VGA circuit  44  includes a second amplifier  146 ; a variable value capacitor  147  in parallel with second amplifier  146 ; a second switch  148  alternating between open and closed states in accordance with a clock φ 2  in parallel with second amplifier  146 ; and a fixed value input capacitor connected to the output of first amplifier  136 . The third stage  133  of the CDS/VGA circuit  44  includes a third amplifier  156 ; a variable value capacitor  157  in parallel with third amplifier  156 ; a third switch  158  alternating between open and closed states in accordance with a clock φ 1  in parallel with third amplifier  156 ; and a fixed value input capacitor  159  connected to the output of second amplifier  146 . The total gain of the CDSVGA circuit  44  according to the present invention is A=(C 2 /C 3 )*(C 4 /C 5 ) and is adjustable according to the present invention by varying C 3  and C 5 . CDS/VGA circuit  44  according to the present invention uses a two phase non-overlapping clock to perform the indicated CDS functions. The two phase clock according to the present invention also allows image signals to be passed to the output while maintaining a positive polarity signal. First stage  131  performs correlated double sampling (CDS) as follows. When clock φ 1  is high, the feed-through level is sampled across first capacitor C 1 , and the output of the first stage is forced to a predetermined reference voltage level. When clock φ 1  falls, the output voltage Vo 1  of first amplifier  136  follows the input. Second stage  132  operates similarly, except that its switch is controlled by the second phase of the two phase non-overlapping clock. This adds a half clock delay, which is effective to maintain a positive output voltage with respect to the reference level. Third stage  133  operates similarly, but adds another half clock delay. 
   Referring now to  FIG. 5B , there is shown a circuit diagram of an amplifier  146  according to an embodiment of the present invention, which is subject to power down performance during a preview mode of operation. According to one embodiment of the present invention, amplifiers  146  and  136  and  156  are constructed of a similar circuit architecture. The design of the power-down amplifier  146  according to the present invention is symmetrical, permitting the amount of current driven by amplifier  146  to be switched between first and second levels. According to one embodiment of the present invention, the first current level is one-half of the current level of the second current level. Amplifier  146  includes a transistor  170  connected in parallel with series transistors  172  and  171 . Transistor  170  is controlled by a power-down signal, so that during power-down, current which would otherwise pass through series transistors  171  and  172  is instead diverted to by-pass the series transistors  171 ,  172 . The series transistors  171  and  172  are connected in parallel with series transistors  173  and  174 , in a current mirror arrangement which insures that the current flowing through the second set of series transistors  173 ,  174  is a function of the current through the first set of series transistors  171 ,  172 . As a result, if current does not flow through series transistors  171 ,  172  as a result, for example, of the current having been by-passed to flow through transistor  170  during power-down operation, there will consequently also be no current flow through transistors  173 ,  174 . In addition to diverting current, transistor  170  also acts to pull the ibias voltage to a low state or close to ground. The specific functional relationship between series transistors  171 ,  172  and series transistors  173 ,  174  is a linear relationship, according to one embodiment, and even more specifically, the current magnitude through transistors  173 ,  174  will be a factor of four times the current through transistors  171 ,  172 . This is a consequence of the current mirror relationship between the respective transistors which results from the size or w/L ratio of the mirrored devices  173  and  174  to the size or the w/L ratio of the devices being mirrored  171  and  172 . Amplifier  146  further includes a transistor  175  in parallel with diode-connected series transistors  176 - 178 . The series connection between diode-connected series transistors  176 - 178  establishes a long device which is subject to being current by-passed, when transistor  175  is activated during power-down. Transistor  175  acts to provide extra power down functionality for any trickle of current that might have been passed through to transistors  173  and  174 . Simply stated, during power-down operation, any remaining current is drawn through transistor  175 , having the consequence that no current will flow during power-down through transistors  176 - 178 . According to the present invention, the bias circuitry includes transistors  170 - 187  and  270 - 287 ; the amplifier section of the circuitry includes transistors  190 - 193  and  290 - 293 . Transistors  185 - 187  are connected in a current mirror configuration to ensure that the current flowing through the second set of transistors  185 - 187  is a function of the current flowing through the first set of transistors  176 - 178 . The specific functional relationship between series transistors  176 - 178  and series transistors  185 - 187  is linear, according to one embodiment. More specifically, the current flowing through transistors  185 - 187  is equal to the current through transistors  176 - 178 , according to one embodiment. Amplifier  146  further includes a transistor  180  in parallel with the diode-connected series transistors  181 - 183 . This is similar to the series transistors  176 - 178  connected in parallel to transistor  175  with nmos devices instead of pmos devices. Transistor  180  acts to provide extra power down functionality for any trickle of current that might have passed through to transistors  185 - 187 . The operation of transistors  270 - 287  is analogous, except the current input comes from ibias 2  rather than ibias 1 , and the power down signal is PD 2  rather than PD 1 . Transistors  190 - 193  and  290 - 293  operate as amplifier circuitry. In particular, transistors  190  and  290 , have their gates tied to a predetermined voltage level vbias 3 . These devices act as current sources with their current dependent on the voltage level of vbias 3 . Transistors  193  and  293  have their gates tied to Vin and are used to control the output (Vout) with a predetermined transfer characteristic from the input to the output. Transistors  191 ,  192 ,  291 , and  292  have a dual function. During normal operation (non-power down), these transistors are provided with a bias voltage that is set up by the transistors surrounding the amplifier circuitry. In particular, transistors  176 - 178  set up a bias voltage (vbias 2   a ) that goes to the gate of transistor  191 , and transistors  276 - 278  set up the bias voltage (vbias 1   b ) that goes to the gate of transistor  291 . Further, transistors  181 - 183  set up the bias voltage (vbias 1   a ) that goes to the gate of transistor  192 , and transistors  281 - 283  set up the bias voltage (vbias 1   b ) that goes to the gate of transistor  292 . During normal operation, transistors  191 ,  192 ,  291 , and  292  are used as cascode devices, and the bias voltages vbias 1   a,b  and vbias 2   a,b  are set such that these transistors operate in a saturated state. In saturation, these transistors cause an increase in DC gain from the input voltage Vin to the output voltage Vout. Moreover, during normal operation, transistors  192  and  292  act to isolate the input from the output, to eliminate capacitive coupling otherwise present between input and output nodes. During power-down action, transistors  191 ,  192 ,  291 , and  292  turn off the current from respective amplifier circuitry branches. In particular, vbias 2   a,b  nodes are held high, i.e., close to vdd) and vbias 1   a,b  are held low, i.e., close to ground. With these voltages operating on the gates of transistors  191 ,  192 ,  291 , and  292 , the respective transistors act as open switches and do not allow any current to flow through the corresponding amplifier branches. Accordingly, the amplifier  146  is configured to have two independent power down control nodes, respectively PD 1  and PD 2 . Thus, the amplifier  146  can be completely powered down or partially powered down by turning off one or the other of its two symmetrical sides. With half of the amplifier powered down, there is a power savings of one half normal operating power subject to a reduced drive level and a corresponding reduced settling time for amplified signals. During preview operation, the reduced settling time is acceptable, because the resolution needed for video display during preview is reduced and minor settling errors are tolerable. 
   Referring now to  FIG. 6 , there is shown a timing diagram of the operation of a correlated double sampling variable gain amplifier (CDS/VGA)  44  operating with a two-phase clock according to an embodiment of the present invention. In particular, there is shown a timing diagram of the two-phase clock of CDS/VGA circuit  114  and the imager signal, and the output signals of the first, second, and third stages, respectively  131 ,  132 , and  133 . In particular, the falling edge of φ 1  occurs just before the transition from feed-through to active video, for example v( 1 ), of the CCD input signal. The falling edge of clock φ 2  occurs just before the transition from active video to reset of the CCD input signal. Clocks φ 2  and φ 1  are non-overlapping clocks. The output of stage  1  follows the CCD input, when φ1 is low. The output of stage two follows the output of stage  1 , a half clock cycle earlier in time, when φ2 is low. The output of stage  3  follows the output of stage  2  from a half clock cycle earlier in time when φ1 was low. 
   Referring now to  FIG. 7A , there is shown a block diagram of an analog-to-digital converter (ADC)  46  according to one embodiment of the present invention. In particular, the ADC  46  is a 10-bit pipelined ADC which includes nine ADC stages respectively  61 - 69 , of which the last four stages  66 - 69  are turned off during the preview mode of operation in accordance with the present invention. Accordingly, the data from the ADC  46  during preview mode is of reduced resolution—a reduction, however, which is not apparent to the viewer of LCD panel  20 , because the resolution level of LCD panel  20  is inherently hardware limited to a lower level, for example commonly about 6-bits. 
   Referring now to  FIG. 7B , there is shown a diagram according to the present invention, which shows different levels of resolution output from ADC  46 , depending upon whether low significant value stages of ADC  46  are engaged for operation or disengaged. The diagram particularly expresses the relationship between stages of ADC  46  and the output bits from the ADC  46 . Each stage of ADC  46  outputs 2 bits. The two bits output by stg 8  have a bit significance of LSB  1  and LSB. Each other stage&#39;s output has a significance that is twice the value of the subsequent stage&#39;s output. In equation form, this is understood as: stgx_output=two_bit_output*2 (8-x) . The output bits of ADC  46  are thus found by adding the outputs of all of the stages together, with their proper significance. When in preview mode, stg 5 -stg 8  are powered down, and their outputs go to “00”. Thus, bits b 3 -b 0  are always “0000” in preview mode and accordingly contain no information. Thus, the additional resolution which would be provided by stages  5 - 8  is suppressed, as it would not have been relied upon in the expression of information on the face of LCD panel  20 . By turning off the indicated stages of ADC  46 , considerable power and battery savings are made, resulting in improved performance system-wide. 
   In summary according to the present invention, a processing system for a charge coupled device (CCD) or CMOS imaging system includes a correlated double sample (CDS) circuit for receiving data from an imager, a variable gain amplifier (VGA) having amplifiers of selectable current level to enable reduced data resolution in a preview display, a low power mode analog-to-digital converter (ADC) having a selectable narrow bit-width output and coupled to said VGA circuit, and a gain adjust circuit coupled to said ADC. The single chip low-power analog front end produces digitized CCD data in either 13-bit, 12-bit or 10-bit formats at a first current level and 9-bit, 8-bit, or 6-bit formats at a second current level. The VGA amplifier includes symmetrical subcircuits which are independently actuable to enable full or reduced data resolution levels respectively for still image capture operation and video previewing on a separate preview screen.