Patent Publication Number: US-6661729-B2

Title: Semiconductor device having test mode

Description:
This application is a divisional of application Ser. No. 09/930,174 filed Aug. 16, 2001 now U.S. Pat 6,498,760. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a semiconductor device and, more particularly, a semiconductor device having a test mode driven by an external power supply potential. 
     2. Description of the Background Art 
     Conventionally, in a dynamic random access memory (hereinbelow, abbreviated as DRAM), packing density is being increased and power supply voltage is being decreased. For this purpose, the DRAM is provided with an internal power supply potential generating circuit for generating an internal power supply potential by dropping the external power supply potential. In the DRAM, to reject an early defective piece in which a defect occurs relatively early after shipment, a burn-in test is carried out before shipment. In the burn-in test, an internal power supply potential higher than that in a normal mode is applied, and data is written/read to/from each of memory cells under high-temperature environment. It accelerates the occurrence of a defective, so that an early defective piece can be prevented from being shipped. 
     FIG. 10 is a block diagram showing the configuration of such an internal power supply potential generating circuit  80  of a DRAM. In FIG. 10, the internal power supply potential generating circuit  80  includes a VPP generating circuit  81 , a VDDS generating circuit  82 , and a VDDP generating circuit  83 . 
     The VPP generating circuit  81  includes, as shown in FIG. 11, a ring oscillator  84 , a charge pump circuit  85 , and a detector  86 . The detector  86  receives both a potential VPP of a power supply node N 85  and an internal power supply potential VDDS from the VDDS generating circuit  82 . When VPP&lt;VDDS+2 Vthn (where Vthn denotes a threshold voltage of an N-channel MOS transistor), the detector  86  sets a signal φE to the “H” level. When VPP≧VDDS+2 Vthn, the detector  86  sets the signal φE to the “L” level. When the signal φE is at the “H” level, the ring oscillator  84  generates a clock signal CLK and supplies it to the charge pump circuit  85 . When the signal φE is at the “L” level, the ring oscillator  84  is made inactive. The charge pump circuit  85  supplies a predetermined amount of positive charges to the power supply node N 85  in response to the rising edge of the clock signal CLK. 
     When VPP&lt;VDDS+2 Vthn, the positive charges are supplied from the charge pump circuit  85  to the power supply node N 85 . When VPP≧VDDS+2 Vthn, the supply of power from the charge pump circuit  85  to the power supply node N 85  is stopped. The potential VPP at the power supply node N 85  is therefore maintained at VDDS+2 Vthn. The internal power supply potential VPP is used as a wordline selection level. 
     The VDDS generating circuit  82  includes, as shown in FIG. 12, an operational amplifier  90 , a constant current source  91 , a variable resistive element  92 , and P-channel MOS transistors  93  and  94 . The constant current source  91  and the variable resistive element  92  are connected in series between a line of the external power supply potential VCC and a line of a ground potential VSS. The P-channel MOS transistor  93  has the source for receiving an external reference potential VRS′, the drain connected to a node N 91  between the constant current source  91  and the variable resistive element  92 , and the gate for receiving a test signal/TE. 
     The P-channel MOS transistor  94  is connected between the line of the external power supply potential VCC and a power supply node N 94 . The operational amplifier  90  has an inversion input terminal connected to the node N 91 , a non-inversion input terminal connected to the power supply node N 94 , and the output terminal connected to the gate of the P-channel MOS transistor  94 . The operational amplifier  90  and the P-channel MOS transistor  94  construct a voltage follower for maintaining the potential VDDS at the power supply node N 94  at the same level as the potential at the node N 91 . The internal power supply potential VDDS is applied to a sense amplifier. 
     At the time of tuning, the test signal/TE is set to the “H” level as an inactivate level, and the P-channel MOS transistor  93  is made non-conductive. The resistance value of the variable resistive element  92  is tuned so that the internal power supply potential VDDS becomes equal to a predetermined value VRS. 
     At the time of a burn-in test, the test signal/TE is set to the “H” level as an activate level, the P-channel MOS transistor  93  is made conductive, and the internal power supply potential VDDS becomes equal to the external reference potential VRS′(&gt;VRS). The internal power supply potential VPPS becomes equal to VRS′+2 Vthn. In normal operation, the test signal/TE is set to the “H” level as an inactivate level, the P-channel MOS transistor  93  is made non-conductive, and the internal power supply potential VDDS becomes VRS. The internal power supply potential VPP becomes equal to VRS+2 Vthn. 
     The VDDP generating circuit  83  includes, as shown in FIG. 13, an operational amplifier  95 , a constant current source  96 , a variable resistive element  97 , P-channel MOS transistors  98  and  99 , an N-channel MOS transistor  100 , and an inverter  101 . The constant current source  96  and the variable resistive element  97  are connected in series between the line of the external power supply potential VCC and the line of the ground potential VSS. The P-channel MOS transistor  99  is connected between the line of the external power supply potential VCC and a power supply node N 98 . The operational amplifier  95  has an inversion input terminal connected to a node N 96  between the constant current source  96  and the variable resistive element  97 , a non-inversion input terminal connected to the power supply node N 98 , and an output terminal connected to the gate of the P-channel MOS transistor  99 . The operational amplifier  95  and the P-channel MOS transistor  99  construct a voltage follower which maintains the potential VDDP of the power supply node N 98  to the level same as the potential of the node N 96 . The internal power supply potential VDDP is supplied to peripheral circuits. 
     The P-channel MOS transistor  98  is connected in parallel with the constant current source  96 . The N-channel MOS transistor  100  is connected between the gate of the P-channel MOS transistor  99  and the line of the ground potential VSS. The test signal/TE is directly supplied to the gate of the P-channel MOS transistor  98  and also to the gate of the N-channel MOS transistor  100  via the inverter  101 . 
     At the time of tuning, the test signal/TE is set to the “H” level as an inactivate level, and the MOS transistors  98  and  100  are made non-conductive. The resistance value of the variable resistive element  97  is tuned so that the internal power supply potential VDDP becomes equal to a predetermined value VRP (&gt;VRS). 
     At the time of a burn-in test, the test signal/TE is set to the “L” level as an activate level, the MOS transistors  98  and  100  are made conductive, and the internal power supply potential VDDP becomes equal to the external power supply potential VCC. In normal operation, the test signal /TE is set to the “H” level as an inactivate level, the MOS transistors  98  and  100  are made non-conductive, and the internal power supply potential VDDP becomes VRP. 
     In short, in normal operation, VPP=VRS+2 Vthn, VDDS=VRS, and VDDP=VRP. At the time of the burn-in test, VPP=VRS′+2 Vthn, VDDS=VRS′, and VDDP=VCC. VRS and VRP are tuned. 
     In the conventional internal power supply potential generating circuit  80 , however, VPP is equal to VDDS+2 Vthn. Consequently, VPP and VDDS cannot be set independently of each other. Occurrence of an early defective in a circuit portion to which VPP is applied and that in a circuit portion to which VDDS is applied cannot be separately accelerated, so that test efficiency is low. 
     The resistance values of the two variable resistive elements  92  and  97  have to be tuned. The tuning is, however, troublesome. 
     SUMMARY OF THE INVENTION 
     It is, therefore, an object of the invention to provide a semiconductor device having high test efficiency. 
     Another object of the invention is to provide a semiconductor device capable of easily adjusting an internal reference potential. 
     A semiconductor device according to the invention includes: a first reference potential generating circuit of which output potential is adjustable, for outputting a first internal reference potential which is lower than the external power supply potential; a first power supply circuit for maintaining a first power supply node at the first internal reference potential in a normal operation mode, and maintaining the first power supply node at an external reference potential in a test mode; a second power supply circuit for maintaining a second power supply node at a boosted potential higher than the first internal reference potential by a predetermined first voltage in the normal operation mode, and supplying the external power supply potential to the second power supply node in the test mode; a level shifting circuit for outputting a potential obtained by level-shifting the potential of the first power supply node by a predetermined second voltage to the external power supply potential side; a third power supply circuit for maintaining a third power supply node at an output potential of the level shifting circuit; and an internal circuit for receiving a drive power from the first to third power supply circuits via the first to third power supply nodes and performing a predetermined operation. Consequently, in the test mode, the first power supply node is maintained at the first external reference potential and the second power supply node is maintained at the external power supply potential, so that occurrence of a defect in the circuit portion to which the potential of the first power supply node is applied and that in the circuit portion to which the potential of the second power supply node is applied can be accelerated separately from each other. Thus, the test efficiency is increased. Since it is sufficient to adjust only the first internal reference potential, as compared with the conventional technique in which two internal reference potentials have to be adjusted, the internal reference potential can be easily adjusted. 
     Preferably, the first reference potential generating circuit includes: a first constant current source connected between a line of the external power supply potential and a first output node, for supplying a predetermined first current to the first output node; and a first variable resistive element of which resistance value is adjustable, which is connected between the first output node and a line of a ground potential. In this case, by adjusting the resistance value of the first variable resistive element, the first internal reference potential can be adjusted. 
     Preferably, the second power supply circuit includes: a charge pump circuit which is activated when a potential of the second power supply node is lower than the boosted potential in the normal operation mode and supplies a current to the first power supply node; and a switching element which is connected between a line of the external power supply potential and the second power supply node and is made conductive in the test mode. In this case, the second power supply circuit can be easily constructed. 
     Preferably, the level shifting circuit includes: a second constant current source which is connected between the line of the external power supply potential and a second output node and supplies a predetermined second current to the second output node; and a transistor connected between the second output node and the line of the ground potential, of which input electrode receives the potential of the first power supply node. In this case, the predetermined second voltage is used as a threshold voltage of the transistor. 
     Preferably, there is provided a second reference potential generating circuit of which output potential is adjustable, for outputting a second internal reference potential which lies between the external power supply potential and the first internal reference potential, and the third power supply circuit maintains the third power supply node at the second internal reference potential in the normal operation mode, and maintains the third power supply node at an output potential of the level shifting circuit in the test mode. In this case, the potential of the third power supply node in the normal operation mode can be finely adjusted, so that the internal circuit is allowed to operate with high precision. 
     Preferably, the semiconductor device further includes: a second reference potential generating circuit of which output potential is adjustable, for outputting a second internal reference potential which lies between the external power supply potential and the first internal reference potential; and a selecting circuit for selecting either an output potential of the level shifting circuit or the second internal reference potential, and the third power supply circuit maintains the third power supply node at a potential selected by the selecting circuit. In this case, when the output potential of the level shifting circuit is selected, the internal reference potential can be easily adjusted but the precision of the operation of the internal circuit deteriorates. On the other hand, when the second reference potential is selected, the adjustment of the internal reference potential becomes troublesome, but the internal circuit is allowed to operate with high precision. 
     Preferably, the second reference potential generating circuit includes: a third constant current source connected between a line of the external power supply potential and a third output node, for supplying a predetermined third current to the third output node; and a second variable resistive element connected between the third output node and the line of the ground potential, of which resistance value is adjustable. In this case, by adjusting the resistance value of the second variable resistive element, the second internal reference potential can be adjusted. 
     Preferably, the semiconductor device is a semiconductor memory device, and a sense amplifier receives a drive power from the first power supply circuit via the first power supply node, the wordline selected by the row selecting circuit receives a drive power from the second power supply circuit via the second power supply node, and the row selecting circuit, the column selecting circuit, and the write/read circuit receive a drive power from the third power supply circuit via the third power supply node. The present invention is particularly effective on this case. 
     A semiconductor device according to another aspect of the invention includes: a first power supply circuit for generating a first internal power supply potential lower than the external power supply potential; a level shifting circuit for outputting a potential obtained by level-shifting the first internal power supply potential by a predetermined voltage to the external power supply potential side; a second power supply circuit for maintaining a second internal power supply potential at the same level as a predetermined reference potential in a normal operation mode, and maintaining the second internal power supply potential at the same level as an output potential of the level shifting circuit in a test mode; and an internal circuit which is driven by first and second internal power supply potentials generated by the first and second power supply circuits and performs a predetermined operation. In the test mode, the second internal power supply potential is maintained at a potential obtained by level-shifting the first internal power supply potential only by a predetermined voltage, so that the level of the second internal power supply potential can be easily set, and the test efficiency is increased. In the normal operation mode, since the second internal power supply potential is maintained at the predetermined reference potential, the internal circuit is allowed to operate with high precision. 
     Preferably, the predetermined reference potential is a potential between the external power supply potential and the first internal power supply potential. In this case, the reference potential can easily be generated. 
     Preferably, the semiconductor device according to the present invention further includes a reference potential generating circuit of which output potential is adjustable, for outputting the predetermined reference potential. The reference potential generating circuit includes: a first constant current source connected between a line of the external power supply potential and a first output node, for supplying a predetermined first current to the first output node; and a variable resistive element connected between the first output node and a line of a ground potential, of which resistance value is adjustable. In this case, by adjusting the resistance value of the variable resistive element, the potential can be adjusted to the reference potential. 
     Preferably, the level shifting circuit includes: a second constant current source connected between a line of the external power supply potential and a second output node, for supplying a predetermined second current to the second output node; and a first transistor connected between the second output node and the line of the ground potential, of which input electrode receives the first internal power supply potential. In this case, the predetermined voltage is a threshold voltage of the first transistor. 
     Preferably, the second power supply circuit includes: a switching circuit for supplying the reference potential to a third output node in the normal operation mode and supplying an output potential of the level shifting circuit to the third output node in the test mode; a second transistor connected between the line of the external power supply potential and a fourth output node; and a control circuit for controlling an input voltage of the second transistor so that the potentials of the third and fourth output nodes coincide with each other, and the potential of the fourth output node becomes equal to the second internal power supply potential. In this case, the second power supply circuit can be easily constructed. 
    
    
     The foregoing and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings. 
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram showing the configuration of a whole DRAM according to a first embodiment of the invention; 
     FIG. 2 is a circuit block diagram showing the configuration of a memory mat illustrated in FIG. 1; 
     FIG. 3 is a circuit diagram showing the configuration of a sense amplifier and input/output control circuit illustrated in FIG. 2; 
     FIG. 4 is a block diagram showing the configuration of an internal power supply potential generating circuit illustrated in FIG. 1; 
     FIG. 5 is a circuit block diagram showing the configuration of a VPP generating circuit illustrated in FIG. 4; 
     FIG. 6 is a circuit diagram showing the configuration of a VDDP generating circuit illustrated in FIG. 4; 
     FIG. 7 is a circuit diagram showing the configuration of a VDDP generating circuit of a DRAM according to a second embodiment of the invention; 
     FIG. 8 is a circuit diagram showing the configuration of a VDDP generating circuit in a third embodiment of the invention; 
     FIG. 9 is a circuit block diagram showing a modification of the third embodiment; 
     FIG. 10 is a block diagram showing the configuration of an internal power supply potential generating circuit of a conventional DRAM; 
     FIG. 11 is a block diagram showing the configuration of a VPP generating circuit illustrated in FIG. 10; 
     FIG. 12 is a circuit diagram showing the configuration of a VDDS generating circuit illustrated in FIG. 10; and 
     FIG. 13 is a circuit diagram showing the configuration of a VDDP generating circuit illustrated in FIG.  10 . 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     First Embodiment 
     FIG. 1 is a block diagram showing the configuration of a whole DRAM according to a first embodiment of the invention. In FIG. 1, the DRAM includes an internal power supply potential generating circuit  1 , a clock generating circuit  2 , a row and column address buffer  3 , a row decoder  4 , a column decoder  5 , a memory mat  6 , an input buffer  9 , and an output buffer  10 . The memory mat  6  includes a memory array  7  and a sense amplifier and input/output control circuit  8 . 
     The internal power supply potential generating circuit  1  generates internal power supply potentials VPP, VDDS, and VDDP on the basis of a power supply potential VCC, a ground potential VSS, and a reference potential VRS′ which are supplied from the outside, and supplies the generated internal power supply potentials to the whole DRAM. The clock generating circuit  2  selects a predetermined operation mode in accordance with external control signals/RAS and /CAS to control the whole DRAM. 
     The row and column address buffer  3  generates row address signals RA 0  to RAi and column address signals CA 0  to CAi in accordance with external address signals A 0  to Ai (where i denotes an integer of 0 or larger), and supplies the generated signals RA 0  to RAi and CA 0  to CAi to the row decoder  4  and the column decoder  5 , respectively. 
     The memory array  7  includes a plurality of memory cells each for storing one-bit data. Each of the memory cells is disposed in a predetermined address determined by a row address and a column address. 
     The row decoder  4  designates a row address in the memory array  7  in accordance with the row address signals RA 0  to RAi supplied from the row and column address buffer  3 . The column decoder  5  designates a column address in the memory array  7  in accordance with the column address signals CA 0  to CAi supplied from the row and column address buffer  3 . 
     The sense amplifier and input/output control circuit  8  connects the memory cell of the address designated by the row decoder  4  and the column decoder  5  to one end of a data input/output line pair IOP. The other end of the data input/output pair IOP is connected to the input buffer  9  and the output buffer  10 . The input buffer  9  supplies data Dj (where j denotes an integer of 0 or larger) supplied from the outside to the selected memory cell via the data input/output line pair IOP in response to an external control signal/W in a writing mode. The output buffer  10  outputs read data Qj from the selected memory cell to the outside in response to an external control signal/OE in a reading mode. 
     FIG. 2 is a circuit block diagram showing the configuration of the memory array  7  and the sense amplifier and input/output control circuit  8  in the DRAM illustrated in FIG.  1 . FIG. 3 is a circuit diagram specifically showing the configuration of one of columns in the memory array  7  and the sense amplifier and input/output control circuit  8  illustrated in FIG.  2 . 
     Referring to FIGS. 2 and 3, the memory array  7  includes a plurality of memory cells MC arranged in a matrix, wordlines WL provided in correspondence with the rows, and pairs of bit lines BL and /BL provided in correspondence with the columns. Each of the memory cells MC includes an N-channel MOS transistor  32  for access and a capacitor  33  for storing information. The gate of the N-channel MOS transistor  32  in each memory cell MC is connected to the wordline WL of the corresponding row. The N-channel MOS transistor  32  is connected between the bit line BL or /BL of the corresponding column and one (storage node SN) of the electrodes of the capacitor  33  of the memory cell MC. The other electrode of the capacitor  33  of the memory cell MC receives a cell plate potential VCP. One end of each wordline WL is connected to the row decoder  4 . 
     The sense amplifier and input/output control circuit  8  includes a column selection line CSL provided corresponding to each column, a column selection gate  11 , a sense amplifier  12 , an equalizer  13 , a driver  14 , and a pair (IOP) of data input/output lines IO and /IO. The column selection gate  11  includes N-channel MOS transistors  21  and  22  connected between the bit lines BL and /BL and the data input/output lines IO and /IO. The gates of the N-channel MOS transistors  21  and  22  are connected to the column decoder  5  via the column selection line CSL. When the column selection line CSL is raised to the “H” level of the selection level by the column decoder  5 , the N-channel MOS transistors  21  and  22  are made conductive, and the pair of bit lines BL and /BL and the pair of data input/output lines IO and /IO are coupled. 
     The sense amplifier  12  includes N-channel MOS transistors  23  and  24  connected between the bit lines BL and /BL and a node N 12 , and P-channel MOS transistors  25  and  26  connected between the bit lines BL and /BL and a node N 12 ′. The gates of the MOS transistors  23  and  25  are connected to the bit line/BL, and the gates of the MOS transistors  24  and  26  are connected to the bit line BL. The driver  14  includes an N-channel MOS transistor  27  connected between the node N 12  and the line of the ground potential VSS and a P-channel MOS transistor  28  connected between the node N 12 ′ and the line of the internal power supply potential VDDS. The gates of the MOS transistors  27  and  28  receive sense amplifier activate signals SE and /SE, respectively. When the sense amplifier activate signals SE and /SE go high and low, respectively, the MOS transistors  27  and  28  are made conductive, the nodes N 12  and N 12 ′ have the ground potential VSS and the internal power supply potential VDDS, respectively, and the sense amplifier  12  amplifies a very small potential difference between the pair of bit lines BL and /BL to the internal power supply voltage VDDS. 
     The equalizer  13  includes an N-channel MOS transistor  29  connected between the bit lines BL and /BL and N-channel MOS transistors  30  and  31  connected between the bit lines BL and /BL and a node N 13 ′. The gates of the N-channel MOS transistors  29  to  31  are connected to a node N 13 . The node N 13  receives a bit line equalize signal BLEQ, and the node N 13 ′ receives a bit line potential VBL (=VDDS/2). When the bit line equalize signal BLEQ becomes at the “H” level as an activate level, the equalizer  13  equalizes the potentials of the bit lines BL and /BL to the bit line potential VBL. 
     The operation of the DRAM shown in FIGS. 1 to  3  will now be described. In a write mode, the column selection line CSL of the column according to the column address signals CA 0  to CAi is raised to the “H” level of the selection level by the column decoder  5 , and the column selection gate  11  of the column is made conductive. 
     The input buffer  9  supplies the write data Dj given from the outside to the pair of bit lines BL and /BL in the selected column via the data input/output line pair IOP in response to the signal/W. The write data Dj is given as a potential difference between the bit lines BL and /BL. Subsequently, the wordline WL of the row corresponding to the row address signals RA 0  to RAi is raised to the “H” level (internal power supply potential VPP) as the selection level by the row decoder  4 , and the MOS transistor  32  in the memory cell MC of the row is made conductive. In the capacitor  33  in the selected memory cell MC, charges according to the potential of the bit line BL or /BL are accumulated. 
     In a read mode, first, the bit line equalize signal BLEQ is decreased to the “L” level, the N-channel MOS transistors  29  to  31  in the equalizer  13  are made non-conductive, and the operation of equalizing the bit lines BL and /BL is stopped. Subsequently, the wordline WL in the row corresponding to the row address signals RA 0  to RAi is raised to the “H” level as the selection level by the row decoder  4 . In response to this, the potentials of the bit lines BL and /BL change only by a very small amount in accordance with the charge amount of the capacitor  33  in the activated memory cell MC. 
     The sense amplifier activate signals SE and /SE go high and low, respectively, and the sense amplifier  12  is activated. When the potential of the bit line BL is higher than that of the bit line/BL only by a small amount, the resistance values of the MOS transistors  24  and  25  become lower than those of the MOS transistors  23  and  26 , the potential of the bit line BL is raised to the “H” level (internal power supply potential VDDS), and the potential of the bit line/BL is lowered to the “L” level (ground potential VSS). On the contrary, when the potential of the bit line/BL is higher than that of the bit line BL only by a very small amount, the resistance values of the MOS transistors  23  and  26  become lower than those of the MOS transistors  24  and  25 , the potential of the bit line/BL is raised to the “H” level, and the potential of the bit line BL is lowered to the “L” level. 
     The column selection line CSL of the column corresponding to the column address signals CA 0  to CAi is raised to the “H” level as the selection level by the column decoder  5 , and the column selection gate  11  of the column is made conductive. Data of the pair of bit lines BL and /BL of the selected column is given to the output buffer  10  via the column selection gate  11  and the pair of data input/output lines IO and /IO. The output buffer  10  outputs the read data Qj to the outside in response to the signal /OE. 
     The internal power supply potential generating circuit  1  as a feature of the invention will be described in detail hereinbelow. The internal power supply potential generating circuit  1  includes, as shown in FIG. 4, a VPP generating circuit  41 , a VDDS generating circuit  42 , and a VDDP generating circuit  43 . 
     The VPP generating circuit  41  includes, as shown in FIG. 5, a ring oscillator  44 , a charge pump circuit  45 , a detector  46 , an N-channel MOS transistor  47 , an inverter  48 , and an AND gate  49 . The ring oscillator  44  is activated when an output signal φ 49  of the AND gate  49  goes high, generates a clock signal CLK, and supplies the clock signal CLK to the charge pump circuit  45 . The charge pump circuit  45  is driven by the clock signal CLK and supplies a predetermined amount of positive charges to a power supply node N 47  in response to the rising edge of the clock signal CLK. 
     The N-channel MOS transistor  47  is connected between the line of the external power supply potential VCC and the power supply node N 47 . The test signal/TE is supplied to the gate of the N-channel MOS transistor  47  via the inverter  48 . The detector  46  receives both the potential VPP of the power supply node N 47  and the internal power supply potential VDDS generated by the VDDS generating circuit  42 . When VPP&lt;VDDS+2 Vthn, the detector  46  sets the signal φE to the “H” level. When VPP≧VDDS+2 Vthn, the detector  46  sets the signal φE to the “L” level. The AND gate  49  receives the test signal/TE and the output signal φE of the detector  46 , and supplies the signal φ 49  to the ring oscillator  44 . 
     At the time of a burn-in test, the test signal/TE is set to the “L” level, the output signal φ 49  of the AND gate  49  is fixed to the “L” level, the ring oscillator  44  is made inactive, and the driving of the charge pump circuit  45  is stopped. The N-channel MOS transistor  47  is made conductive, and the potential VPP of the power supply node N 47  is equalized to the external power supply potential VCC. 
     In a normal operation, the test signal/TE is set to the “H” level, the output signal φE of the detector  46  passes through the AND gate  49  and becomes the signal φ 49 , and the N-channel MOS transistor  47  is made non-conductive. When the potential VPP of the power supply node N 47  is lower than VDDS+2 Vthn, the signals φE and φ 49  go high, the ring oscillator  44  is made active, and positive charges are supplied from the charge pump circuit  45  to the power supply node N 47 . When the potential VPP of the power supply node N 47  becomes equal to or higher than VDDS+2 Vthn, the signals φE and φ 49  go low, the ring oscillator  44  is made inactive, and the supply of positive charges from the charge pump circuit  45  to the power supply node N 47  is stopped. The potential VPP of the power supply node N 47  is therefore maintained at VDDS+2 Vthn. The potential VPP of the power supply node N 47  is supplied to the selected wordline WL. VPP is set to be equal to VDDS+2Vthn for the reason that, by suppressing a voltage drop of the N-channel MOS transistor  32  in the memory cell MC, a sufficiently high potential is applied to the storage node SN. 
     Referring again to FIG. 4, the VDDS generating circuit  42  has the same configuration as that of the conventional VDDS generating circuit  82  shown in FIG.  12 . At the time of a burn-in test, the internal power supply potential VDDS is maintained at the same level as the external reference potential VRS′. At the time of normal operation, the internal power supply potential VDDS is maintained at the same level as the internal reference potential VRS. The internal power supply potential VDDS is supplied to the sense amplifier  12  via the driver  14  and also to the VDDP generating circuit  43 . 
     The VDDP generating circuit  43  includes, as shown in FIG. 6, an operational amplifier  50 , a constant current source  51 , and P-channel MOS transistors  52  and  53 . The constant current source  51  and the P-channel MOS transistor  52  are connected in series between the line of the external power supply potential VCC and the line of the ground potential VSS, and the gate of the P-channel MOS transistor  52  receives the internal power supply potential VDDS from the VDDS generating circuit  42 . The potential of the source (node N 51 ) of the P-channel MOS transistor  52  is equal to VDDS+Vthp (where Vthp denotes a threshold voltage of the P-channel MOS transistor). The P-channel MOS transistor  53  is connected between the line of the external power supply potential VCC and the power supply node N 53 . The operational amplifier  50  has an inversion input terminal connected to the node N 51 , the non-inversion input terminal connected to the node N 53 , and an output terminal connected to the gate of the P-channel MOS transistor  53 . The operational amplifier  50  and the P-channel MOS transistor  53  construct a voltage follower which maintains the potential VDDP of the power supply node N 53  at the same level as that of the potential VDDS+Vthp of the node N 51 . 
     Therefore, at the time of a burn-in test, the internal power supply potential VDDP becomes VRS′+Vthp. At the time of a normal operation, the internal power supply potential VDDP becomes equal to VRS+Vthp. The internal power supply potential VDDP is supplied to peripheral circuits such as the clock generating circuit  2  and the row and column address buffer  3 . VDDP is set to be larger than VDDS to increase the operation speed of the peripheral circuit and for the reason that since the withstand voltage of a transistor in the peripheral circuit is set to be higher than that of a transistor in the sense amplifier  12  or the like, even when a high voltage is applied to the peripheral circuit, no problem occurs. 
     In short, in normal operation, VPP=VRS+2 Vthn, VDDS=VRS, and VDDP=VRS+Vthp. At the time of the burn-in test, VPP=VCC, VDDS=VRS′, and VDDP=VRS′+Vthp, and VRS is tuned. 
     In the first embodiment, at the time of the burn-in test, VPP=VCC and VDDS=VRS′. Consequently, VPP and VDDP can be set independently of each other. Occurrence of an early defective in a circuit portion to which VPP is applied and that in a circuit portion to which VDDS is applied can be separately accelerated, so that test efficiency is high. 
     Since it is sufficient to tune only the resistance value of the variable resistive element  92 , as compared with the conventional technique in which the resistance values of the two variable resistive elements  92  and  97  have to be tuned, the work for the tuning is reduced. 
     Second Embodiment 
     In the first embodiment, by setting VDDP=VDDS+Vthp, the work of tuning is lessened. Since the access speed of the DRAM is determined by the internal power supply potential VDDP, when the highly accurate access speed is required, it is desirable to tune the internal power supply potential VDDP. The second embodiment solves this problem. 
     FIG. 7 is a circuit diagram showing the configuration of a VDDP generating circuit  60  of a DRAM according to the second embodiment of the invention. Referring to FIG. 7, the VDDP generating circuit  60  is different from the VDDP generating circuit  43  in FIG. 6 with respect to the point that a constant current source  61 , a variable resistive element  62 , a P-channel MOS transistor  63 , an N-channel MOS transistor  64 , and an inverter  65  are added. 
     The constant current source  61  and the variable resistive element  62  are connected in series between the line of the external power supply potential VCC and the line of the ground potential VSS. The P-channel MOS transistor  63  is connected between the node N 61  positioned between the constant current source  61  and the variable resistive element  62  and the inversion input terminal of the operational amplifier  50 . The N-channel MOS transistor  64  is connected between the node N 51  and the inversion input terminal of the operational amplifier  50 . The test signal/TE is supplied to the gates of the MOS transistors  63  and  64  via the inverter  65 . 
     At the time of tuning, the test signal/TE is set to the “H” level as the inactivate level, the P-channel MOS transistor  63  is made conductive, the N-channel MOS transistor  64  is made non-conductive, and the potential of the node N 61  is applied to the inversion input terminal of the operational amplifier  50 . The resistance value of the variable resistive element  62  is tuned so that the internal power supply potential VDDP becomes equal to the predetermined value VRP. 
     At the time of the burn-in test, the test signal/TE is set to the “L” level as the activate level, the P-channel MOS transistor  63  is made non-conductive, the N-channel MOS transistor  64  is made conductive, and the potential VDDS+Vthp=VRS′+Vthp of the node N 51  is applied to the inversion input terminal of the operational amplifier  50 . Therefore, the internal power supply potential VDDP becomes equal to VRS′+Vthp. 
     In normal operation, the test signal/TE is set to the “H” level as the inactivate level, the P-channel MOS transistor is made conductive, the N-channel MOS transistor  64  is made non-conductive, and the potential of the node N 61  is applied to the inversion input terminal of the operational amplifier  50 . The internal power supply potential VDDP therefore becomes equal to VRP. Since the other configuration and operation are the same as those of the first embodiment, their description will not be repeated. 
     In the second embodiment, in the normal operation, the tuned potential VDDP=VRP is supplied to the peripheral circuit. Consequently, the access speed can be set with high precision. 
     Third Embodiment 
     FIG. 8 is a circuit diagram showing the configuration of a VDDP generating circuit  70  of a DRAM according to a third embodiment of the invention. Referring to FIG. 8, the VDDP generating circuit  70  is different from the VDDP generating circuit  60  in FIG. 7 with respect to the point that the inverter  65  is eliminated, and a change-over switch  71 , a P-channel MOS transistor  72 , an N-channel MOS transistor  73 , an OR gate  74 , and an inverter  75  are added. 
     The gates of the MOS transistors  63  and  64  are connected to a common terminal  71   c  of the change-over switch  71 . One switch terminal  71   a  and the other switch terminal  71   b  of the change-over switch  71  receive the external power supply potential VCC and the ground potential VSS, respectively. The change-over switch  71  is switched by, for example, connection of a bonding wire, replacement of a contact mask, and the like. FIG. 8 shows a state where the terminals  71   a  and  71   c  are made conductive. 
     The P-channel MOS transistor  72  is connected in parallel with the constant current source  61 . The N-channel MOS transistor  73  is connected between the gate of the P-channel MOS transistor  53  and the line of the ground potential VSS. The OR gate  74  receives both the test signal/TE and a signal φC appearing at the common terminal  71   c  of the change-over switch  71 , and an output signal of the OR gate  74  is directly supplied to the gate of the P-channel MOS transistor  72  and also to the gate of the N-channel MOS transistor  73  via the inverter  75 . 
     When the DRAM is not shipped as a model required to have high-precision access speed, the terminals  71   a  and  71   c  of the change-over switch  71  are connected. Consequently, the signal φC goes high, the N-channel MOS transistor  64  is made conductive, and the P-channel MOS transistor  63  is made non-conductive, so that the VDDP generating circuit  70  has the same configuration as that of the VDDP generating circuit  43  in FIG.  6 . In this case, therefore, the same effects as those of the first embodiment can be produced. 
     When the DRAM is of a model required to have high-precision access speed, the terminals  71   b  and  71   c  of the change-over switch  71  are connected. Consequently, the signal φC goes low, the P-channel MOS transistor  63  is made conductive, and the N-channel MOS transistor  64  is made non-conductive, so that the VDDP generating circuit  70  comes to have the same configuration as that of the conventional VDDP generating circuit  83  in FIG.  13 . In this case, therefore, the access speed can be set with high precision. Since the other configuration is the same as that of the second embodiment, the description will not be repeated. 
     In the third embodiment, the VPP generating circuit  41  may be replaced by the VPP generating circuit  76  in FIG.  9 . The VPP generating circuit  76  is different from the VPP generating circuit  41  in FIG. 5 with respect to the point that the inverter  48  is replaced by the inverter  77  and an NOR gate  78 . The test signal/TE is supplied to one of input nodes of the NOR gate  78 , the signal φC is supplied to the other input node of the NOR gate  78  via the inverter  77 , and an output signal of the NOR gate  78  is supplied to the gate of the N-channel MOS transistor  47 . 
     In the case where the signal φC is at the “H” level, the VPP generating circuit  76  has the same configuration as the VPP generating circuit  41  in FIG.  5 . In the case where the signal φC is at the “L” level, the N-channel MOS transistor  47  is fixed to the non-conductive state, and the VPP generating circuit  76  has the same configuration as the conventional VPP generating circuit  80  in FIG.  11 . 
     Although the present invention has been described and illustrated in detail, it is clearly understood that the same is by way of illustration and example only and is not to be taken by way of limitation, the spirit and scope of the present invention being limited only by the terms of the appended claims.