Patent Publication Number: US-6667702-B2

Title: Semiconductor device having DACs and method of testing such semiconductor device

Description:
FIELD OF THE INVENTION 
     The invention relates to a semiconductor device having a digital-to-analog converters (DACs) for use in digital TV set, DVDs, and game machines, and more particularly to such semiconductor device having DACs suitable for testing the semiconductor device. The invention also relates to a method of testing a semiconductor device. 
     BACKGROUND OF THE INVENTION 
     In digital TV sets, DVDs, and game machines, data processing of a video signal for example is first performed in digital form and then the digital data is converted to an analog data prior to outputting the data. To perform the conversion, a DAC is used. Of course such DAC is integrated in an integrated circuit, so that a dedicated tester is used to assess its performance. 
     FIG. 1 is a block diagram representation of a conventional semiconductor  100  having a DAC. As shown in FIG. 1, the semiconductor device  100  is provided with n-bit digital input signal Din and a clock CLK. The input signal Din is latched in a latch circuit  101 , decoded in a decoder  102 , and then converted into an analog signal by a DAC  103 . The converted analog signal is output from the DAC as the analog output signal Dout. 
     In testing the semiconductor device  100 , it is connected to a tester  110 . The tester  110  provides to the semiconductor device  100  a digital input signal Din and a clock CLK which are identical to the signals actually supplied to the semiconductor device  100 . The output analog signal Dout is assessed by the tester. 
     When the semiconductor device  100  is designed for a high-frequency (e.g. 135 MHz) video signal, the performance of the DAC in the high-frequency range must be verified. In general, distortions of the output analog signal Dout are measured for a given digital input signal Din to analyze various characteristics of the DAC  103 . To do this, the tester  110  provides a digital sinusoidal wave signal and a clock to the semiconductor device  100 , and distortions of the analog signal output from the semiconductor device  100  are measured by a spectrum analyzer set up in the tester  110 . 
     Thus, in testing the semiconductor device having a DAC, a tester capable of outputting high frequency signals in different test modes is necessary. That is, the high-frequency tester  110  must be capable of realizing various problematic conditions that it might encounter in the measurements of high frequency signals as output from the semiconductor device, for example variation in capacitance of wires, signal delays in the wires, and timing violation in the semiconductor. Because of this requirement, the tester is considerably expensive as compared with ordinary mass-produced testers (e.g. testers designed to output a signal of 40 MHz). Consequently, a capital investment for the testers greatly affects the manufacturing cost of the semiconductor device having a DAC. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of the invention to provide a semiconductor device having DACs that can be successfully tested in various high-frequency modes without using a tester capable of outputting high-frequency digital test signals. 
     It is another object of the invention to provide a method of testing a semiconductor device having such DACs. 
     In accordance with one aspect of the invention, there is provided a semiconductor device equipped with at least one channel of DAC for converting a digital signal input to the semiconductor device into an analog signal before it is output from the semiconductor device, the semiconductor device comprising: 
     test pattern generation means for storing and generating test patterns; and 
     a clock input terminal which can receive a test clock during a test, wherein 
     the test pattern generation means generates a digital test signal representing a test pattern based on a test clock supplied to the clock input terminal during a test, and provides the digital test signal and the test clock to the DAC channel. 
     In testing the semiconductor device equipped with such DAC according to the invention, the device only require a fast test clock, which can be easily generated and supplied from an external clock generator. Based on this test clock, a high-frequency digital test signal representing a prescribed test pattern can be generated by the test pattern generation means and provided to the DAC channel, so that the DAC can be tested for the high-frequency signal as in the test using a high-frequency tester. 
     Therefore, although the semiconductor device requires therein such pattern generation means, it can be provided at low cost since no expensive tester generating a high frequency test signal is required to test the device. 
     The semiconductor device may further comprise a test mode input terminal for receiving a test mode signal indicative of a test mode supplied to the test pattern generation means so that, based on the test clock received through the test clock terminal during a test, the test pattern generation means generates a digital test signal representing the test pattern designated by the test mode signal. 
     In this way, many of the characteristics of the DAC can be tested using different test patterns in accord with various test modes. 
     The semiconductor device may comprise an additional test power supply terminal for supplying a voltage higher than the operating voltage of the DAC to the test pattern generation means. This will improve the performance of the test pattern generation means and allow it to operate at a higher speed. Thus, it is possible to minimize the size of the test pattern generation means while driving it faster, which helps minimization of the chip size of the semiconductor device. 
     The semiconductor device shown herein has six channels of DACs (referred to as DAC channels) such that the three DACs in the first three DAC channels and the three DACs in the second three DAC channels can be separately tested. The first three channels include a composite signal channel, a brightness channel, and a chromatic signal channel, while the second three channels include a brightness signal channel, a first color difference signal channel, and a second color difference signal channel. Hence, six channels can be tested using three testing elements by switching the first and the second three channels. 
     The semiconductor device has a multiplicity of DAC channels such that the DACs in the respective channels can be individually tested. This can be done by turning off the DACs (thereby causing the DACs to generate outputs at an intermediate level), except one DAC. This arrangement enables measurement of cross talks between the active (ON) channel and rest of the inactive (OFF) channels. 
     The test pattern generation means may store sinusoidal wave test patterns to generate a sinusoidal wave test pattern in accord with a designated test mode when said test mode is a sinusoidal wave test mode. These sinusoidal wave test patterns have different combinations of frequencies, amplitudes, and DC biases in combination. In some of the test modes, the sinusoidal wave test pattern has an arbitrary number of the lower bits fixed to zero. 
     In such test mode, the level and/or the distortion of a high-frequency signal can be tested. Different frequencies, levels (in amplitude/accuracy) and DC biases may be used in combination to carry out varied sinusoidal wave tests. In addition, bit-drop by the DAC may be tested by evaluating the level of the analog signal output therefrom, based on the fact that the level of the output analog signal varies if the input digital signal is missing certain bits. 
     The test pattern generation means may store square wave test patterns to generate a square wave test pattern in accord with a designated test mode when the test mode is a square wave test mode. These square wave test patterns have different periods, duty cycles, amplitudes, and DC biases in combination. The amplitude of the square wave pattern may have an arbitrary number of lower bits fixed to zero. 
     In these square wave test modes, it is possible to evaluate glitch energy and the transient response of the DAC at the rise/fall of the signal. Various tests may be also carried out using different square wave test patterns having different periods, duty cycles, levels (amplitudes/precisions), and DC biases. In addition, bit-drop by the DAC may be tested by evaluating the level of the analog signal output therefrom, based on the fact that the level of the output analog signal varies if the input digital signal is missing certain bits. 
     The DAC may be a current additive type DAC which comprises: 
     a set of multiple conversion units connected in parallel, each unit having a constant current element controlled by a reference potential and a switching element connected in series with said constant current element and adapted to be turned ON/OFF by a control signal based on the input digital data; 
     reference potential generation means for generating said reference potential; 
     a reference potential line for providing said reference potential to each of said constant current elements of said conversion units, wherein 
     the reference potential generation means has a potential feedback means for setting low its impedance of the reference generation means as viewed from the reference potential line. 
     In this arrangement of the current additive type DAC, the impedance of the reference potential generation means is set low as viewed from the reference potential line so that the cutoff frequency as determined by the floating static capacitance and the impedance of the reference potential generation means is shifted to a higher frequency, so that the variation of the reference potential on the reference potential line is reduced, resulting in a high fidelity D/A conversion. 
     In accordance with the invention, there is provided a method of testing a semiconductor device having DACs for converting digital signals into analog signals, utilizing a tester having a slower processing speed than the DACs, wherein the tester has means for generating a test clock; the semiconductor device has test pattern generation means for storing and generating test patterns for testing at least a portion of the performance of the DACs, and a clock input terminal which can receive the test clock to drive the test pattern generation means by the test clock, said method comprising steps of: 
     supplying a test clock from the tester to the test pattern generation means via said clock input terminal; 
     causing the test pattern generation means to generate a digital test signal in synchronism with the test clock; 
     supplying the test clock and the digital test signal to the DACs to convert the digital test signal into an analog signal; and 
     assessing the analog signal by the tester. 
     The method of the invention allows tests of the DACs at high frequency without using an expensive tester that generates high-frequency digital test signals. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram representation of test system for testing a conventional semiconductor device having a DAC. 
     FIG. 2 is a schematic circuit diagram of a semiconductor device having DAC channels according to the invention, and a tester. 
     FIG. 3 is a circuit diagram of preferred DAC channels according to the invention. 
     FIG. 4 is a circuit diagram shown for explanations of the DAC of FIG.  3 . 
     FIG. 5 is the circuit diagram of another preferred DAC according to the invention. 
     FIG. 6 shows a test pattern for use in a sinusoidal wave test. 
     FIG. 7 shows another test pattern for use in another sinusoidal wave test. 
     FIG. 8 shows a test pattern for use in a square wave test. 
     FIG. 9 shows another test pattern for use in another square wave test. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 2 schematically shows the structure of a semiconductor device  10  having DAC channels according to the invention, along with a tester  40 . In this figure, the semiconductor device  10  is assumed to have six DAC channels. 
     We first describe the structure of the semiconductor device  10 . As shown in FIG. 2, an n-bit composite digital signal Nin includes all the signals necessary for a color picture such as a brightness signal Y, a chromatic signal for orthogonal phase modification of a first and a second color difference signals B-Y and R-Y, respectively, which are modulated by a sub-transmission wave (which has 3.58 MHz in NTSC system or 4.43 MHz in PAL system), and horizontal/vertical synchronizing signals. The number of bits, n, can be 10 for example. It is assumed that other digital signals have the same number of bits, unless otherwise stated. Other portions of the circuit not relevant to the inventive DAC channels are omitted in FIG.  2 . 
     A first digital brightness signal Yin 1  includes horizontal and vertical synchronizing signals. Digital chromatic signal Cin includes a first color difference signal B-Y, and a second color difference signal R-Y. 
     Of these 3-channel signals, the first digital brightness signal Yin 1  and the digital chromatic signal Cin constitutes a set of color signals. Since the set of color signals separately includes the first digital brightness signal Yin 1  and digital chromatic signal Cin, Y/C separation is not needed to obtain a color image of good quality. The digital composite signal Nin alone includes another set of color signals. 
     A second digital brightness signal Yin 2  is the same as the first digital brightness signal Yin 1 . Digital first color difference signal Uin is a B-Y signal, while digital second color difference signal Vin is an R-Y signal. The 3-channel signals consisting of the second digital brightness signal Yin 2 , a first digital color difference signal Uin, and a second digital color difference signal Vin, together form a set of color signals. Since the brightness signal and the respective color difference signals are separately formed in the set, no Y/C separation nor separation of the chromatic signal is needed. Hence, color pictures of exceedingly high-fidelity may be obtained. 
     These 6-channel digital signals, that is, the composite signal Nin, brightness signal Yin 1 , chromatic signal Cin, brightness signal Yin 2 , first color difference signal Uin, and second color difference signal Vin, are formed from a set of RGB color signals. They are synchronized to the same clock. System clock CLK is provided to a first set of latch circuits  11  (latch circuits  11 - 1 - 11 - 3 ), a first set of decoders  12  (decoders  12 - 1 - 12 - 3 ), a first set of DACs  13  (DACs  13 - 1 - 13 - 3 ), a second set of latch circuits  21  (latch circuits  21 - 1 - 21 - 3 ), a second set of decoders  22  (decoders  22 - 1 - 22 - 3 ), and a second set of DACs  23  (DACs  23 - 1 - 23 - 3 ). The 6-channel digital signals and the system clock CLK supplied to the semiconductor device  10  are high frequency signals of 135 MHz, for example. 
     The first three of the 6-channel digital signals are latched by the respective latch circuits  11 - 1 - 11 - 3 . The latched digital signals are decoded by the respective decoders  12 - 1 - 12 - 3  and then converted into analog signals by the respective DACs  13 - 1 - 13 - 3 . They are output as analog composite signal Nout, first analog brightness signal Yout 1 , and analog chromatic signal Cout, respectively. It is noted that each of the decoders  12 - 1 - 12 - 3  is adapted to decode an n-bit digital signal into predetermined m-bit codes adequate for the DACs  13 - 1 - 13 - 3 , so that the decoders  12 - 1 - 12 - 3  may not be necessary for a certain-type of the DACs  13 - 1 - 13 - 3 . This is also the case in other channels. 
     Similarly, in the second three channels, digital signals are latched by the respective latch circuits  21 - 1 - 21 - 3 , decoded by the respective decoders  22 - 1 - 22 - 3 , and converted to analog signals by the respective DACs  23 - 1 - 23 - 3 . They are output as a set of second analog brightness signal Yout 2 , analog first color difference signal Uout, and analog second color difference signal Vout. 
     In place of the set of color signals consisting of the second digital brightness signal Yin 2 , digital first color difference signal Uin, and digital second color difference signal Vin, another set of color signals Rin, Gin, and Bin may be used. 
     The DACs  13 - 1 - 13 - 3  and  23 - 1 - 23 - 3  of the invention have a circuit structure that is adequate for high speed operation to deal with high frequency signals. FIGS. 3 and 5 illustrate preferred current additive type DACs. FIG. 4 illustrates a current additive type DAC which is useful in understanding the operation of the DACs shown in FIGS. 3 and 5. 
     The DAC of FIG. 3 has conversion units  51 - 5   n  for converting coding signals into analog current signals. Each of the conversion units  51 - 5   n  is provided with a respective p-type MOS FET (hereinafter referred to as p-type transistor)  511 - 5   n   1  serving as a constant current element and a respective p-type transistor  512 - 5   n   2  serving as a switching element. The gates of the p-type transistors  511 - 5   n   1  are connected with a reference potential line  70  which is in turn coupled with a reference potential generation means  80 . 
     Receiving an input n-bit (e.g. 10 bits) digital signal Din, a decoder  90  decodes it into predetermined m-bit codes, which are applied to the respective gates of the p-type transistors  512 - 5   n   2  as ON/OFF signals. 
     These conversion units  51 - 5   n  are connected in parallel and provide constant currents i 1 -in as determined by the codes supplied thereto by the decoder  90 . These currents are added together to a current in the range of 0 to several 10 mA before it is supplied to a resistive load  60 . 
     The reference potential generation means  80  has a structure as described below. A first p-type transistor  81  is connected in series with a constant current power circuit  82  which is configured to provide a reference current Iref of 100 μA for example. They are connected between a source potential Vdd and the ground potential Gnd. Connected between the gate of the first p-type transistor  81  and the ground potential Gnd is a second p-type transistor  83 . The gate of the second p-type transistor  83  is connected to the drain of the first p-type transistor  81 . Connected between the gate of the first p-type transistor  81  and the source potential Vdd is a resistor  84 . 
     In addition, connected in parallel with the first p-type transistor  81  is a third p-type transistor  87  which is driven by a control signal CONT to make the current additive type DAC operable or inoperable. The p-type transistor  87  is turned ON or OFF in accordance with the potential level of the control signal CONT applied to the gate of the transistor  87 . The second p-type transistor  83  is configured such that a current of about 1 mA flows through it when the current additive type DAC is in operation. 
     Because of this reference potential generation means  80 , the third p-type transistor  87  is turned ON and the second p-type transistor  83  is turned OFF when the control signal CONT is pulled down to a low level (LOW), points B and A (and hence the reference potential line  70 ) have a potential close to the source potential Vdd, thereby rendering the current additive type DAC inoperable. Since the second p-type transistor  83  is turned OFF under this inoperative condition of the DAC, no wasteful current flows through the transistor  83 . 
     When the control signal CONT is pulled up to a high level (HIGH), the third p-type transistor  87  is turned OFF to make the current additive type DAC operable, thereby bringing potentials at points B and A (and hence the potential of the reference potential line  70 ) to the reference potential. This reference potentials is applied via the reference potential line  70  to the gates of the respective p-type transistors  511 - 5   n   1  serving as constant current-controlled elements of the respective conversion units  51 - 5   n.    
     Particularly in the DAC shown in FIG. 3, the second p-type transistor  83  and the resistor  84  together function as a potential feed back means for setting low the impedance of the reference potential generation means  80  as viewed from the reference potential line  70 . 
     Accordingly, the influence of potential variation on the reference potential line  70  via floating capacitances Cp 1 -Cpn caused by the switching of the respective conversion units  51 - 5   n  is reduced. As a result, variations in the gate voltages of the p-type transistors  511 - 5   n   1  of the conversion units  51 - 5   n  become small. Hence, provision of the output current i 1 -in by the respective conversion units  51 - 5   n  is stabilized, thereby permitting high-quality D/A conversion. 
     Referring to FIG. 4 which illustrates a typical DAC for better understanding of the invention, operation of the inventive DAC shown in FIG. 3 will now be described. Like reference numerals denote like or corresponding elements in the two Figures. 
     The DAC shown in FIG. 4 differs from the DAC of the invention shown in FIG. 3 in that the former has a reference potential generation means  80 A, which includes a p-type transistor  81  and a constant current source  82  such that the gate and drain of the transistor  81  are connected together. The rest of the elements of the DAC  80 A are the same as the corresponding elements of the DAC of the invention. 
     As will be understood from FIG. 4, the current Ids passing through the transistor  81  is given by 
     
       
           Ids=k ( Va−Vt ) 2   (1) 
       
     
     where k is a constant parameter having a value, Va is voltage between the source and the gate (hereinafter referred to as source-gate voltage) of the transistor  81 , and Vt is threshold voltage. 
     To evaluate the AC component of the current, Equation (1) is integrated. The result is 
     
       
           ∂Ids/∂Va= 2 k ( Va−Vt )  (2) 
       
     
     so that the impedance Za at point A is given by 
     
       
           Za=∂Va/∂Ids= 1/{2 k ( Va−Vt )}  (3). 
       
     
     For example, if k=625×10 −6 , Va=1.0 (V), and Vt=0.6 (V), then the impedance Za=2 (KΩ). 
     Assuming that the parasitic capacitance Cp (Cp 1 -Cpn) is 5 (pF), the cutoff frequency Fc defined by Fc=1/(2πZa Cp) of the high-pass filter formed of the parasitic capacitance Cp and the impedance Za turns out to be about 16 MHz. 
     With this cutoff frequency Fc (=16 MHz), DACs for video signals for example which undergo high frequency switching (typically at 135 MHz) will be affected by the floating capacitance Cp. 
     In the invention, the DAC as shown in FIG. 3 lowers the impedance Z of the reference potential generation means  80  as viewed from the reference potential line  70  so that the cutoff frequency Fc determined by the floating capacitance Cp and the impedance Z is shifted to a higher frequency. 
     The first and the second p-type transistors  81  and  83  of FIG. 3 may be the same as the p-type transistor  81  of FIG.  4 . The constant current source  82  may be the same as the corresponding constant current source  82  of FIG.  4 . Then, given a source-gate voltage Vb of 1.865 (V), currents Ids 1  and Ids 2  passing through the first and the second p-type transistors  81  and  83 , respectively, turns out to be:                    Ids1   =         k        (     Va   -   Vt     )       2     =     625   ×     10     -   6       ×       (     1.0   -   0.6     )     2                     =     100                   (   µA   )                     (   4   )                     Ids2   =         k        (     Vb   -   Vt     )       2     =     625   ×     10     -   6       ×       (     1.865   -   0.6     )     2                     =     1                     (   mA   )     .                     (   5   )                         
     The mutual conductance Gm of the first p-type transistor  81  is 
     
       
           Gm=∂Ids 1 /∂Va= 2 k ( Va−Vt )=0.5×10 −3 . 
       
     
     The mutual conductance Gm has a negative feedback in such a way that if the potential at point A lowers, Ids 1  of the first p-type transistor  81  increases to raise the potential at point B, which in turn raises the potential at point A by the source follower action of the second p-type transistor  83 . Thus, if the impedance of the constant current source  82  is 100 KΩ for example, the feedback gain equals 100×10 3 ×Gm or about 50. 
     The impedance Z 83  of the second p-type transistor  83  is given by 
     
       
           Z 83 =∂Vb/∂Ids 2=1/{2 k ( Vb−Vt )}=632(Ω)  (3). 
       
     
     Under the feedback effect, the impedance Za at point A is then 
     
       
         632(Ω)×1/50=12.64(Ω). 
       
     
     This impedance Za at point A is the impedance Z of the reference potential generation means  80  as viewed from the reference potential line  70 . Thus, if the parasitic capacitance Cp (Cp 1 -Cpn) is assumed to be 5 (pF) as of the DAC shown in FIG. 4, the cutoff frequency Fc of the high-pass filter made up of the parasitic capacitance Cp and the impedance Z is given by 
     
       
           Fc= 1/(2π Z Cp ), 
       
     
     which is about 2.5 GHz. 
     Since this cutoff frequency Fc (=2.5 GHz) is sufficiently higher than the frequency of Video signals in the neighborhood of 135 MHz, the DAC will not be affected by the floating capacitance Cp associated with the switching of the conversion units. 
     Furthermore, since the reference potential generation means  80  is provided with the third p-type transistor  87  which is turned ON/OFF in accordance with the operative and inoperative status of the DAC, power consumed by the DAC is reduced when the DAC is not in operation. 
     FIG. 5 shows a preferred current additive type DAC in the form of a semiconductor integrated circuit (IC) according to the invention. The DAC shown in FIG. 5 has the same structure as the one shown in FIG. 3 except that the reference potential generation means  80 B differs from the corresponding potential generation means  80 . The rest of the elements are the same in both Figures. 
     The reference potential generation means  80 B is formed as follows. A first p-type transistor  81  and a constant current source  82  providing a reference current Iref (which is 100 μA for example) are connected in series between the source potential Vdd and the ground potential Gnd. A resistor  84  and a second p-type transistor  83  are connected in series between the source potential Vdd and the ground potential Gnd, with the gate of the second p-type transistor  83  connected with the drain of the first p-type transistor  81 . Connected between the gate of the first p-type transistor  81  and the power supply voltage Vdd is an n-type MOS FET (hereinafter referred to as n-type transistor)  85 , with the gate of the n-type transistor  85  connected with the source of the second p-type transistor  83 . Connected between the gate of the first p-type transistor  81  and the ground potential Gnd is a resistor  86 . 
     In addition, as in the example shown in FIG. 3, a third p-type transistor  87  driven by a control signal CONT is connected in parallel with the first p-type transistor  81  to control the operation of the current additive DAC. 
     The DAC of FIG. 5 has a feature that it can be used with a lower voltage power source. On the other hand, in the DAC of FIG. 3, the impedance Z of the reference potential generation means  80  is set low so as to shift the cutoff frequency Fc to a high frequency, based on the fact that Fc is determined by the floating capacitance Cp and the impedance Z. However, this configuration is not suited to a case where a low voltage power supply is used, since the source-gate voltage Va of the first p-type transistor  81  plus the source-gate voltage Vb of the second p-type transistor  83  is applied in series, making the potential at point B as low as Vdd−Va−Vb=Vdd−2.865 (V). 
     In contrast, in the DAC of FIG. 5, the potential at point B becomes Vdd−Va+Vc−Vb, where Va, Vb, and Vc are the source-gate voltage of the first p-type transistor  81 , second p-type transistor  83 , and the n-type transistor  85 , respectively. Therefore, if the circuit parameters are adjusted to make Vc equal to Vb, the potential at point B becomes Vdd−Va, which is sufficiently high even when a low voltage power supply is used. 
     Incidentally, it would be understood that although the DAC shown in FIG. 5 works through the n-type transistor  85 , its operation is essentially the same as the DAC of FIG. 3, and further details of the operation will not be repeated here. 
     It would be noted that in either of the current additive DACs of FIGS. 3 and 5, the impedance of the reference potential generation means  80 / 80 B as viewed from the reference potential line  70  is set low, so that the cutoff frequency Fc determined by the floating capacitance Cp and the impedance of the reference potential generation means  80 / 80 B is shifted to a higher value, thereby reducing the variation of the potential of the reference potential line  70  and allowing highly accurate D/A conversion. 
     Referring back to FIG. 2 again, the test pattern generation circuit  31  is shown to include a ROM  32  as a storage device, and a control section CTR 33 . The ROM  32  stores various n-bit (e.g. 10-bit) test pattern data. Under the control of the control section CTR  33 , a particular test pattern stored in the ROM  32  is retrieved in the form of digital data. The retrieved test pattern is supplied as a digital input signal to the digital input terminals and to the respective latches  11 - 1 - 21 - 3  along with a test clock CLKt. The test clock CLKt has the same frequency of 135 MHz as the system clock CLK. It is noted that the impedance of the test pattern generation circuit  31  is set high (Hi-Z) unless it is in test operation that the output from test pattern generation circuit  31  will not affect the digital input signal during a normal operation. 
     The test patterns stored in the ROM  32  may be categorized into sinusoidal wave patterns and square wave patterns. 
     The sinusoidal wave patterns include many combinations of various frequencies, levels (in amplitude and accuracy), and DC biases. The square wave patterns include many combinations of various periods, duty ratios, levels (in amplitude and accuracy), and DC biases. It should be understood that the levels (in amplitude and accuracy) implies here various amplitudes of the test digital signals and accuracies of the amplitudes available in the test. 
     The test pattern generation circuit  31  is supplied with a test mode signal from the tester  40  via a test mode designation terminal provided in the semiconductor device  10 . The test mode specifies which one of the test patterns stored in the ROM  32  is to be generated in the test. The test clock CLKt having the same frequency of 135 MHz as the system clock CLK is also supplied from the tester  40  to the test pattern generation circuit  31  via the test clock input terminal. 
     It is noted that the test pattern generation circuit  31  is provided with a test power supply terminal for receiving a test voltage Vt independently of the proper power supply voltage Vdd (not shown) which for the semiconductor device  10 . For example, latches  11 - 1 - 11 - 3 , decoders  12 - 1 - 12 - 3 , and DACs  13 - 1 - 13 - 3  utilize the nominal voltage of the power supply voltage Vdd (which is for example 2.5 V). On the other hand, the test power supply voltage Vt supplied to the test pattern generation circuit  31  may be higher (e.g. 4 V) than the nominal voltage (2.5 V) within a permissible voltage range. 
     Such a higher test power supply voltage Vt than the nominal voltage of the semiconductor device  10  improves operability of each circuit element and increases the operating speed thereof. Accordingly, the dimensions of the test pattern generation circuit  31  may be minimized while increasing its operating speed, so that the size of the entire semiconductor chip need not be increased by the test pattern generation circuit  31 . 
     It is noted that it does not matter if it is used under a severe condition than normal operating condition, since the test pattern generation circuit  31  is used only in the test of the semiconductor device  10  unlike other circuit elements. 
     Next, the tester  40  will now be described. The tester  40  need not a fast digital circuit except for a clock generator SG  43 . That is, it can work with a semiconductor device  10  if it is capable of outputting slower digital data than the semiconductor device  10 . In this sense the tester  40  is different from conventional ones. 
     The test mode designation section  41  is adapted to instruct the test pattern generation circuit  31  which test mode is to be used in the test stored in the test pattern generation circuit  31 . The test pattern generation circuit  31  then generates the test pattern-required in the designated mode. At the same time the designated test mode is informed to a test evaluation means provided in the tester  40  for evaluating the test. 
     The test power supply  42  generates test power supply voltage Vt, which is supplied to the test pattern generation circuit  31 . The test power supply voltage Vt (4 Volts for example) is higher than the nominal operating voltage (2.5 V) of the semiconductor device  10 . The SG  43  comprises an oscillator generating a test clock CLKt having the same frequency 135 MHz as that of the system clock CLK. The test clock CLKt is provided to the test pattern generation circuit  31 . 
     The test evaluation means comprises an analog-to-digital converter (referred to as ADC), a spectrum analyzer  45 , and an audio analyzer  46 . 
     The ADC  44  is adapted to convert analog test data received from the semiconductor device  10  into a fast (e.g. 20 MHz) digital data to provide measured data in digital form. The spectrum analyzer  45  also analyzes the analog signal indicative of the test result by measuring the spectrum of the data, which is suited for evaluating the distortion of a sinusoidal wave. The audio analyzer  46  also performs frequency analysis of the analog test data to obtain frequency distribution thereof, which is suited for the evaluation of low-frequency components of a signal (e.g. below 100 KHz). 
     Based on the evaluations made by the ADC  44 , spectrum analyzer  45 , and audio analyzer  46 , a pass-fail determination is made on the first and the second sets of DACs  13 - 1 - 13 - 3 ,  23 - 1 - 23 - 3 . It may be appreciated that a high-frequency digital signal is not involved in the tester  40  so that timing of the digital elements is simple. 
     A process of testing the semiconductor device  10  having such a DAC as described above by using the tester  40  will now be described. 
     First, a test mode is set in a test mode designation section  41 . The data indicative of the test mode is supplied to the test pattern generation circuit  31 , ADC  44 , spectrum analyzer  45 , and audio analyzer  46  via test mode instruction terminals thereof. The test pattern generation circuit  31  is also supplied with a test clock CLKt from the SG 43  and a test power supply voltage Vt from the power source  42  for the test (referred to as test power source). 
     A digital test signal for the designated test mode generated by the test pattern generation circuit  31  and the test clock CLKt are input to the first set of latch circuits  11 - 1 - 11 - 3  or the second set of latch circuits  21 - 1 - 21 - 3 . The digital signal is converted into analog signal by means of the first set of latch circuits  11 - 1 - 11 - 3 , decoders  12 - 1 - 12 - 3 , and DACs  13 - 1 - 13 - 3 , or by the second set of latch circuits  21 - 1 - 21 - 3 , decoders  22 - 1 - 22 - 3 , and DACs  23 - 1 - 23 - 3 . The analog signal is fed to the evaluation means (i.e. ADC  44 , spectrum analyzer  45 , and audio analyzer  46 ) of the tester  40  for evaluation. Based on the evaluation, pass-fail determinations of the first and the second sets of DACs  13 - 1 - 13 - 3  and  23 - 1 - 23 - 3  are performed. 
     Test modes used in the invention will now be further described in detail. The test modes can be categorized into two groups: a first test mode for simultaneously testing three channels; and a second test mode for testing individual DACs of the three channels independently by turning OFF the DACs except for one. 
     In the first test mode, either three channels of composite signal Nin channel, first brightness signal Yin 1  channel, and chromatic signal Cin channel or of second brightness signal Yin 2  channel, first color difference signal Uin channel, and second color difference signal Vin channel can be simultaneously tested. Thus, all the 6 channels of the two sets of three channels can be tested using three-channel evaluation means by switching off one of the sets. 
     In the second test mode, three channels of one set can be individually turned ON/Off for a test. If the outputs of the DAC channels to be turned OFF are fixed to an intermediate level, cross talks between the operative and inoperative channels may be measured. 
     Each of the first and the second test modes may be further categorized into two groups. They are: (a) sinusoidal wave test modes in which a level and distortion of a high-frequency input signal can be tested, and (b) square wave test modes in which glitch energy and transient response at a rise/fall of wave can be tested. 
     In the sinusoidal wave modes, out of various sinusoidal wave patterns stored in the ROM  32 , a desired sinusoidal wave pattern having in combination a desired frequency, level (amplitude or accuracy), and DC bias is selected. For example, a frequency may be selected from 2.4 MHz (=135 MHz×73/4096), 11.6 MHz (135 MHz×11/128), and 1.55 MHz (135 MHz×1/128). An accuracy mode is provided in which a few lowest bits of 10-bit signal are fixed to zero to test bit dropping by a DAC. In a DC bias test mode, a predetermined DC value is added to a sinusoidal wave. 
     The test patterns for use in these test modes are illustrated in FIGS. 6 and 7. FIG. 6 shows a 10-bit sinusoidal wave pattern having a full scale level of  1023 , with its lowest one, two, and three bit(s) sequentially dropped off in time. Since the level of the output analog signal changes if bit-dropping occurs in the DAC, it is possible to test bit-dropping by the DAC through observation of the output level of the DAC. FIG. 7 shows a pattern in which a sinusoidal wave having its upper 6 bits shifted to the lower bits is level-shifted in sequence by three different DC biases of 0, 256, 512, and 768. In this mode, video signals having a DC component superposed with AC components may be tested. “Steps” on the abscissas of FIGS. 6 and 7 denotes the number of clocks. 
     In these sinusoidal wave modes, a sinusoidal wave having a given frequency of 2.4 MHz (135 MHz×73/4096) for example goes through all the levels from 0 to 10 bits (0-1023) over one or more periods. These modes permits sophisticated or higher-level reliability tests of the DAC. 
     In the square wave modes, out of various square wave patterns stored in the ROM  32 , a desired square wave pattern having in combination a desired frequency, level (in amplitude or accuracy), and DC bias is selected. For example, a duty ratio can be selected from 1/2, 2/4, 1/16, and 15/16. As in sinusoidal wave test mode, an accuracy mode is provided in which a few lowest bits of 10-bit signal fixed to zero to test bit dropping by a DAC. In a DC bias mode, a predetermined DC current is added to a square wave. 
     The test patterns for use in these test modes are illustrated in FIGS. 8 and 9. FIG. 8 shows a 10-bit square wave pattern having a full scale level of  1023 , which sequentially changes its period from 2 Step to 4 and 16 Steps and its duty ratio from 1/2 to 2/4 and 1/16. These changes in the digital signal in the square wave tests are reflected in the output level of the analog signal, from which results of a test can be obtained. 
     FIG. 9 shows a square wave pattern in which a square wave having a period of 2 Steps and duty ratio of 1/2 is generated, which changes its DC bias cyclically from 0 to 256, 512, and 768, and its lowest 1-7 bits varied. This wave pattern can be used to test, for example, a video signal having a DC component superposed with an AC component. Also in FIGS. 8 and 9, “Step” on the respective abscissas denotes the number of clocks. 
     As describe above in detail, various test patterns can be selectively output from the test pattern generation circuit  31  (ROM  32 ) in the form of digital data by the test mode designation section  41  to the internal DACs  13 - 1 - 13 - 3  or  23 - 1 - 23 - 3  together with a clock in accordance with an external test clock CLKt.