Patent Publication Number: US-2023136512-A1

Title: Resonant converter and voltage conversion method

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of International Application No. PCT/CN2020/099052, filed on Jun. 29, 2020, the disclosure of which is hereby incorporated by reference in its entirety. 
    
    
     TECHNICAL FIELD 
     The embodiments relate to the field of electronic circuit technologies, a resonant converter, and a voltage conversion method. 
     BACKGROUND 
     As one type of DC-DC converter, an inductor-inductor-capacitor (LLC) resonant direct current (DC)-DC converter is widely used in the communication and energy field due to high conversion efficiency. However, when an input-output voltage of the LLC resonant DC-DC converter needs to be adjusted in a wide range, conversion efficiency of the LLC resonant DC-DC converter is low. 
     SUMMARY 
     The embodiments may provide a resonant converter and a voltage conversion method, so that the resonant converter implements high conversion efficiency during wide input-output voltage power conversion. 
     A first aspect provides a resonant converter. The resonant converter may include a high-frequency inversion circuit, an LLC resonant tank network, and a hybrid rectification circuit. The LLC resonant tank network is separately coupled to the high-frequency inversion circuit and the hybrid rectification circuit. The high-frequency inversion circuit is configured to convert a first direct current voltage into a first alternating current voltage. The LLC resonant tank network is configured to adjust the first alternating current voltage to obtain a second alternating current voltage. The hybrid rectification circuit works in a full-bridge rectification mode when a direct current voltage adjustment gain falls within a first threshold range, or works in a voltage doubling rectification mode when a direct current voltage adjustment gain falls within a second threshold range, to convert the second alternating current voltage into a second direct current voltage for output, where the direct current voltage adjustment gain is a ratio of the second direct current voltage to the first direct current voltage. 
     In the embodiments, when the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit of the resonant converter may work in the full-bridge rectification mode to convert the second alternating current voltage into the second direct current voltage for output. When the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit of the resonant converter may work in the voltage doubling rectification mode to convert the second alternating current voltage into the second direct current voltage for output. The hybrid rectification circuit is switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     In a possible implementation, the resonant converter further includes a voltage conversion circuit. The voltage conversion circuit is separately coupled to the high-frequency inversion circuit and the LLC resonant tank network. The voltage conversion circuit is configured to adjust the first alternating current voltage to obtain a third alternating current voltage. That the LLC resonant tank network adjusts the first alternating current voltage to obtain a second alternating current voltage includes: The LLC resonant tank network adjusts the third alternating current voltage to obtain the second alternating current voltage. 
     In the embodiments, the resonant converter may further include the voltage conversion circuit. When the voltage conversion circuit performs voltage conversion, a primary side coil and a secondary side coil may be electrically isolated. 
     In a possible implementation, the resonant converter further includes a voltage conversion circuit. The voltage conversion circuit is separately coupled to the LLC resonant tank network and the hybrid rectification circuit. The voltage conversion circuit is configured to adjust the second alternating current voltage to obtain a fourth alternating current voltage. That the hybrid rectification circuit works in a full-bridge rectification mode when a direct current voltage adjustment gain falls within a first threshold range, or works in a voltage doubling rectification mode when a direct current voltage adjustment gain falls within a second threshold range, to convert the second alternating current voltage into a second direct current voltage includes: The hybrid rectification circuit works in the full-bridge rectification mode when the direct current voltage adjustment gain falls within the first threshold range, or works in the voltage doubling rectification mode when the direct current voltage adjustment gain falls within the second threshold range, to convert the fourth alternating current voltage into the second direct current voltage for output. 
     In the embodiments, the resonant converter may further include the voltage conversion circuit. When the voltage conversion circuit performs voltage conversion, a primary side coil and a secondary side coil may be electrically isolated. 
     In a possible implementation, the high-frequency inversion circuit is any high-frequency inversion circuit in a half-bridge inversion circuit, a full-bridge inversion circuit, and a three-level inversion circuit. 
     In a possible implementation, the LLC resonant tank network includes a first capacitor, a first inductor, and a second inductor. 
     In a possible implementation, a first end of the first inductor and a first end of the first capacitor are respectively coupled to two output ends of the high-frequency inversion circuit, a second end of the first inductor and a second end of the first capacitor are respectively coupled to two ends of the second inductor, and the two ends of the second inductor are respectively coupled to two input ends of the hybrid rectification circuit. Alternatively, a first end of the first inductor and a first end of the second inductor are respectively coupled to two output ends of the high-frequency inversion circuit, a second end of the first inductor and a second end of the second inductor are respectively coupled to two ends of the first capacitor, and two ends of the second inductor are respectively coupled to two input ends of the hybrid rectification circuit. Alternatively, a first end of the first capacitor and a first end of the second inductor are respectively coupled to two output ends of the high-frequency inversion circuit, a second end of the first capacitor and a second end of the second inductor are respectively coupled to two ends of the first inductor, and two ends of the second inductor are respectively coupled to two input ends of the hybrid rectification circuit. 
     In a possible implementation, the hybrid rectification circuit includes a second capacitor and a rectifier bridge. A first input end of the rectifier bridge is coupled to a first end of the second capacitor, a second end of the second capacitor is coupled to a first output end of the LLC resonant tank network, a second input end of the rectifier bridge is coupled to a second output end of the LLC resonant tank network, and two output ends of the rectifier bridge are used as output ends of the resonant converter. The rectifier bridge may include a plurality of switching transistors or may include at least one diode and at least one switching transistor. 
     In the embodiments, the hybrid rectification circuit of the resonant converter includes the second capacitor and the rectifier bridge. The second capacitor may implement the following: When the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit works in the voltage doubling rectification mode, to convert the second alternating current voltage into the second direct current voltage for output. It may be understood that only one capacitor needs to be added to the hybrid rectification circuit, and the following can be implemented without adding a bidirectional switching transistor: When the direct current voltage adjustment gain falls within different ranges, the hybrid rectification circuit is switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     In a possible implementation, the voltage conversion circuit includes a third inductor and a transformer. Two ends of the third inductor are respectively coupled to two output ends of the high-frequency inversion circuit, the two ends of the third inductor are respectively coupled to two ends of a primary side of the transformer, and two ends of a secondary side of the transformer are respectively coupled to two input ends of the LLC resonant tank network. Alternatively, two ends of the third inductor are respectively coupled to two output ends of the LLC resonant tank network, the two ends of the third inductor are respectively coupled to two ends of a primary side of the transformer, and two ends of a secondary side of the transformer are respectively coupled to two input ends of the hybrid rectification circuit. 
     A second aspect provides a voltage conversion method. The method is applied to the resonant converter provided in the first aspect, the resonant converter includes a high-frequency inversion circuit, an LLC resonant tank network, and a hybrid rectification circuit, and the method includes: converting a first direct current voltage into a first alternating current voltage by using the high-frequency inversion circuit; adjusting the first alternating current voltage by using the LLC resonant tank network to obtain a second alternating current voltage; and enabling, when a direct current voltage adjustment gain falls within a first threshold range, the hybrid rectification circuit to work in a full-bridge rectification mode to convert the second alternating current voltage into a second direct current voltage for output; or enabling, when a direct current voltage adjustment gain falls within a second threshold range, the hybrid rectification circuit to work in a voltage doubling rectification mode to convert the second alternating current voltage into a second direct current voltage. 
     In the embodiments, when the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit may work in the full-bridge rectification mode to convert the second alternating current voltage into the second direct current voltage for output. When the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit may work in the voltage doubling rectification mode to convert the second alternating current voltage into the second direct current voltage for output. The hybrid rectification circuit is switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     In a possible implementation, the resonant converter further includes a voltage conversion circuit, and the method further includes: adjusting the voltage of the first alternating current by using the voltage conversion circuit to obtain a third alternating current. The adjusting the voltage of the first alternating current by using the LLC resonant tank network to obtain a second alternating current includes: adjusting the voltage of the third alternating current by using the LLC resonant tank network to obtain the second alternating current. 
     In a possible implementation, the resonant converter further includes a voltage conversion circuit, and the method further includes: adjusting the voltage of the second alternating current by using the voltage conversion circuit to obtain a fourth alternating current voltage. The enabling, when a direct current voltage adjustment gain falls within a first threshold range, the hybrid rectification circuit to work in a full-bridge rectification mode to convert the second alternating current voltage into a second direct current voltage for output includes: enabling, when the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit to work in the full-bridge rectification mode to convert the fourth alternating current voltage into the second direct current voltage. The enabling, when a direct current voltage adjustment gain falls within a second threshold range, the hybrid rectification circuit to work in a voltage doubling rectification mode to convert the second alternating current voltage into a second direct current voltage for output includes: enabling, when the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit to work in the voltage doubling rectification mode to convert the fourth alternating current voltage into the second direct current voltage for output. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG.  1    is a schematic diagram of a structure of a transformer according to an embodiment; 
         FIG.  2    is a schematic diagram of a structure of a resonant converter according to an embodiment; 
         FIG.  3    is a schematic diagram of a gain curve according to an embodiment; 
         FIG.  4    is a schematic diagram of a structure of another resonant converter according to an embodiment; 
         FIG.  5    is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  6    is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  7    is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  8    is a schematic diagram of a structure of a rectifier bridge according to an embodiment; 
         FIG.  9    is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  10    is a schematic diagram of a structure of coupling an LLC resonant tank network to a voltage conversion circuit according to an embodiment; 
         FIG.  11    is a diagram of a working principle of a resonant converter according to an embodiment; 
         FIG.  12    is a schematic diagram of another gain curve according to an embodiment; 
         FIG.  13    is a schematic diagram of still another gain curve according to an embodiment; 
         FIG.  14 A  is a schematic diagram of a waveform according to an embodiment; 
         FIG.  14 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  14 C  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  14 D  is a schematic diagram of another waveform according to an embodiment; 
         FIG.  14 E  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  14 F  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  14 G  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  15 A  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  15 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  15 C  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  15 D  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  15 E  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  15 F  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  15 G  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  16 A  is a schematic diagram of still another gain curve according to an embodiment; 
         FIG.  16 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  16 C  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  16 D  is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  16 E  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  16 F  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  16 G  is a schematic diagram of still another waveform according to an embodiment; 
         FIG.  17    is a schematic diagram of a structure of still another resonant converter according to an embodiment; 
         FIG.  18    is a schematic diagram of a structure of another LLC resonant tank network according to an embodiment; and 
         FIG.  19    is a schematic flowchart of a voltage conversion method according to an embodiment. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     The embodiments may provide a resonant converter and a voltage conversion method, so that the resonant converter implements high conversion efficiency during wide input-output voltage power conversion. The following describes the embodiments in detail with reference to the accompanying drawings. It is clear that the described embodiments are merely some, but not all, of the embodiments. 
     To facilitate better understanding of the resonant converter and the voltage conversion method provided in the embodiments, the following first describes an application scenario of the embodiments. A DC-DC converter is a voltage converter that effectively outputs a fixed voltage after converting an input voltage. Generally, the DC-DC converter includes an inversion circuit, a transformer, and a rectification circuit. To enable an output voltage of the DC-DC converter to change in a wide range,  FIG.  1    is a schematic diagram of a structure of a transformer according to an embodiment. As shown in  FIG.  1   , a relay may be connected in series on a secondary-side winding of the transformer, and a turn ratio between a primary-side winding and the secondary-side winding is adjusted by using the relay, to adjust an output voltage. However, the relay cannot implement online fast switching, and a circuit needs to be powered off when the turn ratio is switched. This limits application of the relay. 
     As one type of DC-DC converter, an LLC resonant converter can implement a zero voltage switch (ZVS) of a primary-side switching transistor and a zero current switch (ZCS) of a secondary-side rectifier, and therefore has high conversion efficiency. The LLC resonant converter may adjust a gain of the LLC resonant converter by adjusting a frequency of a switching transistor, so that an output voltage of the LLC resonant converter can be adjusted in a range. Generally, the frequency of the switching transistor is adjusted in a range. Therefore, a change range of the output voltage of the LLC resonant converter is also fixed. When a change range of the frequency of the switching transistor is specified, if the change range of the output voltage is increased, conversion efficiency of the LLC resonant converter is reduced. 
       FIG.  2    is a schematic diagram of a structure of a resonant converter according to an embodiment. As shown in  FIG.  2   , the resonant converter may include an inversion circuit, an LLC resonant tank network, a transformer, and a rectification circuit.  FIG.  3    is a schematic diagram of a gain curve according to an embodiment.  FIG.  3    is a diagram of a gain curve existing when the resonant converter in  FIG.  2    works. As shown in  FIG.  3   , a horizontal coordinate is a frequency of a switching transistor, and a vertical coordinate is a gain of the resonant converter. The frequency of the switching transistor changes between f mid  and f max . When the inversion circuit works in a half-bridge mode, the gain of the resonant converter changes between M min  and M mid . When the inversion circuit works in a full-bridge mode, the gain of the resonant converter changes between M mid  and M max . It may be understood that, when the frequency of the switching transistor changes between f min  and f max , and the inversion circuit works in the half-bridge mode or the full-bridge mode, the resonant converter can change only in a small gain range, and correspondingly, an output voltage can change only in a small range. To increase a change range of the output voltage, the inversion circuit may be switched between the half-bridge mode and the full-bridge mode. When the inversion circuit is switched between the half-bridge mode and the full-bridge mode for working, the gain of the resonant converter may change between M min  and M max , and a change range of the gain is increased, so that the change range of the output voltage can be increased. f min  is a minimum frequency of the switching transistor, f max  is a maximum frequency of the switching transistor, M min  is a minimum gain of the resonant converter, M mid  is a switching-point gain of the resonant converter, and M max  is a maximum gain of the resonant converter. Therefore, the inversion circuit of the resonant converter is switched between the full-bridge mode and the half-bridge mode, so that the resonant converter can have a wider gain range. Therefore, the output voltage of the resonant converter can change in a wide range, so that conversion efficiency of the resonant converter can be improved when a change range of the frequency of the switching transistor is specified. However, when an input voltage is low, and the inversion circuit of the resonant converter works in the half-bridge mode, current stress of the switching transistor is large. In addition, if continuous voltage adjustment needs to be performed for the circuit, a half-bridge LLC resonant tank gain needs to be at least twice a full-bridge LLC resonant tank gain, a ratio of a maximum voltage to a minimum voltage of an LLC resonant tank voltage adjustment gain is greater than 2, a resonant tank gain adjustment range is wide, and a reactive power cycle is large. Consequently, efficiency is not high. 
       FIG.  4    is a schematic diagram of a structure of another resonant converter according to an embodiment. As shown in  FIG.  4   , the resonant converter may include a high-frequency inversion circuit  101 , an LLC resonant tank network  102 , and a hybrid rectification circuit  103 . 
     The LLC resonant tank network  102  may be separately coupled to the high-frequency inversion circuit  101  and the hybrid rectification circuit  103 . 
     The high-frequency inversion circuit  101  is configured to convert a first direct current voltage into a first alternating current voltage. 
     The LLC resonant tank network  102  is configured to adjust the first alternating current voltage to obtain a second alternating current voltage. 
     The hybrid rectification circuit  103  works in a full-bridge rectification mode when a direct current voltage adjustment gain falls within a first threshold range, or works in a voltage doubling rectification mode when a direct current voltage adjustment gain falls within a second threshold range, to convert the second alternating current voltage into a second direct current voltage for output, where the direct current voltage adjustment gain is a ratio of the second direct current voltage to the first direct current voltage. 
     The first direct current voltage is a direct current voltage that is input to the resonant converter and may be denoted as V in . An input end of the high-frequency inversion circuit  101  may be used as an input end of the resonant converter. A direct current voltage may be input by using the input end. The direct current voltage may be provided by a direct current power supply, may be provided by a rectification circuit, or may be provided by another circuit that has a same function. An output end of the hybrid rectification circuit  103  may be used as an output end of the resonant converter. An output voltage of the resonant converter may be denoted as V o . The resonant converter may supply power to a load, the output end thereof may be coupled to the load, and the load may be at least one of a resistor, a capacitor, a circuit, and the like. For example, as shown in  FIG.  5   , the load may include a load capacitor and a load resistor. 
     The hybrid rectification circuit  103  may include two working modes, namely, the full-bridge rectification mode and the voltage doubling rectification mode. The working mode of the hybrid rectification circuit  103  may be switched based on the direct current voltage adjustment gain. When the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit  103  may work in the full-bridge rectification mode. When the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit  103  may work in the voltage doubling rectification mode. The first threshold range may be less than the second threshold range. 
     In an embodiment, the resonant converter may further include a voltage conversion circuit. 
       FIG.  5    is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  5   , in an implementation, the voltage conversion circuit  104  may be separately coupled to the high-frequency inversion circuit  101  and the LLC resonant tank network  102 . The voltage conversion circuit  104  is configured to adjust the first alternating current voltage to obtain a third alternating current voltage. That the LLC resonant tank network adjusts the first alternating current voltage to obtain a second alternating current voltage may be that the LLC resonant tank network  102  adjusts the third alternating current voltage to obtain the second alternating current voltage. 
     The voltage conversion circuit  104  may be located between the high-frequency inversion circuit  101  and the LLC resonant tank network  102  and may convert an output voltage of the high-frequency inversion circuit  101  and then output the voltage to the LLC resonant tank network  102 . 
       FIG.  6    is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  6   , in another implementation, the voltage conversion circuit  104  may be separately coupled to the LLC resonant tank network  102  and the hybrid rectification circuit  103 . The voltage conversion circuit  104  is configured to adjust the second alternating current voltage to obtain a fourth alternating current voltage. That the hybrid rectification circuit  103  works in a full-bridge rectification mode when a direct current voltage adjustment gain falls within a first threshold range, or works in a voltage doubling rectification mode when a direct current voltage adjustment gain falls within a second threshold range, to convert the second alternating current voltage into a second direct current voltage for output may be as follows: The hybrid rectification circuit  103  works in the full-bridge rectification mode when the direct current voltage adjustment gain falls within the first threshold range, or works in the voltage doubling rectification mode when the direct current voltage adjustment gain falls within the second threshold range, to convert the fourth alternating current voltage into the second direct current voltage for output. 
     The voltage conversion circuit  104  may be located between the LLC resonant tank network  102  and the hybrid rectification circuit  103  and may convert an output voltage of the LLC resonant tank network  102  and then output the voltage to the hybrid rectification circuit  103 . 
       FIG.  7    is a schematic diagram of a structure of still another resonant converter according to an embodiment. The resonant converter shown in  FIG.  7    is obtained by optimizing the resonant converter shown in  FIG.  4   . 
     The high-frequency inversion circuit  101  may include a first switching transistor S p1 , a second switching transistor S p2 , a third switching transistor S p3 , and a fourth switching transistor S p4 . A drain of S P1  and a source of S p2  are used as the input end of the resonant converter. The drain of S P1  is coupled to a drain of S p3 . A source of S p1  is separately coupled to a drain of S p2  and a first input end of the voltage conversion circuit  104 . A source of S p3  is separately coupled to a drain of S p4  and a second input end of the voltage conversion circuit  104 . The source of S p2  is coupled to a source of S p4 . The high-frequency inversion circuit  101  shown in  FIG.  7    is a full-bridge inversion circuit. In addition, the high-frequency inversion circuit  101  may be a half-bridge inversion circuit, a three-level inversion circuit, or another inversion circuit that has a same function. A detailed structure is not described herein. 
     In an embodiment, the LLC resonant tank network  102  may include a first inductor L r1 , a second inductor L r2 , and a first capacitor C r . In an implementation, a first end of L r1  and a first end of C r  are respectively coupled to two output ends of the high-frequency inversion circuit  101 , a second end of L r1  and a second end of C r  are respectively coupled to two ends of L r2 , and the two ends of L r2  are respectively coupled to two input ends of the hybrid rectification circuit  103 . In another implementation, a first end of L r1  and a first end of L r2  are respectively coupled to two output ends of the high-frequency inversion circuit  101 , a second end of L r1  and a second end of L r2  are respectively coupled to two ends of C r , and two ends of L r2  are respectively coupled to two input ends of the hybrid rectification circuit  103 . In still another implementation, a first end of C r  and a first end of L r2  are respectively coupled to two output ends of the high-frequency inversion circuit  101 , a second end of C r  and a second end of L r2  are respectively coupled to two ends of L r1  and two ends of L r2  are respectively coupled to two input ends of the hybrid rectification circuit  103 . 
     The hybrid rectification circuit  103  may include a second capacitor C p  and a rectifier bridge. 
     A first input end of the rectifier bridge is coupled to a first end of C p , a second end of C p  is coupled to a first output end of the LLC resonant tank network  102 , a second input end of the rectifier bridge is coupled to a second output end of the LLC resonant tank network  102 , and two output ends of the rectifier bridge are used as the output end of the resonant converter. 
       FIG.  8    is a schematic diagram of a structure of a rectifier bridge according to an embodiment. As shown in  FIG.  8   , the rectifier bridge may include a plurality of switching transistors. In an embodiment, the rectifier bridge may include four switching transistors. The rectifier bridge may also include at least one diode and at least one switch transistor. In an embodiment, the rectifier bridge may include four diodes and one switching transistor, and the switching transistor may be connected in parallel with any diode. For example, a switching transistor S R  may be connected in parallel with a diode D R2 . In another embodiment, the rectifier bridge may include three diodes and one switching transistor. In still another embodiment, the rectifier bridge may include two diodes and two switching transistors. In still another embodiment, the rectifier bridge may include one diode and three switching transistors. The diode may be a parasitic diode of a switching transistor. 
     In an implementation, as shown in  FIG.  8   , the rectifier bridge may include a fifth switching transistor S R1 , a sixth switching transistor S R2 , a seventh switching transistor S R3 , and an eighth switching transistor S R4 . A drain of S R1  is coupled to a drain of S R3 . A source of S R1  is separately coupled to the second output end of the LLC resonant tank network  102  and a drain of S R2 . A source of S R2  is coupled to a source of S R4 . A source of S R3  is separately coupled to the second end of C p  and a drain of S R4 . The drain of S R3  and the source of S R4  are used as the output end of the resonant converter. 
     When the load includes a load capacitor C o  and a load resistor R o , as shown in  FIG.  8   , a first end of C o  may be separately coupled to a first output end of the hybrid rectification circuit  103  and a first end of the load resistor R o , and a second end of C o , a second output end of the hybrid rectification circuit  103 , and a second end of R o  are separately used to connect to a ground terminal. 
       FIG.  9    is a schematic diagram of a structure of still another resonant converter according to an embodiment. The resonant converter shown in  FIG.  9    is obtained by optimizing the resonant converter shown in  FIG.  5   . As shown in  FIG.  9   , the voltage conversion circuit  104  may include a third inductor L m  and a transformer T. 
     Two ends of L m  are respectively coupled to two output ends of the high-frequency inversion circuit  101 , the two ends of L m  are respectively coupled to two ends of a primary side of T, and two ends of a secondary side of T are respectively coupled to two input ends of the LLC resonant tank network  102 . 
     T may be a center tap transformer or may be a common single-output winding transformer. Current directions of the primary side and the secondary side of T are the same. 
       FIG.  10    is a schematic diagram of a structure of coupling an LLC resonant tank network to a voltage conversion circuit according to an embodiment. As shown in  FIG.  10   , in three implementations, the two input ends of the LLC resonant tank network  102  are coupled to two output ends of the voltage conversion circuit  104 . For a coupling manner, refer to descriptions in  FIG.  8   . Details are not described herein again. 
     A gain of the resonant converter is M total =V o /V in =M SP *N SP *M LLC *M SR  where M SP  is a gain of the high-frequency inversion circuit  101 , N SP  is a transformer ratio of the voltage conversion circuit  104 , M LLC  is a gain of the LLC resonant tank network  102 , and M SR  is a gain of the hybrid rectification circuit  103 . N SP  is a fixed value and cannot be adjusted, but M SP , M LLC  and M SR  can be adjusted. In the following specification, M LLC  is adjusted after it is assumed that M SP  and M SR  are fixed values. A value of M SP  varies with a structure of the M SP  high-frequency inversion circuit  101 . For example, if the high-frequency inversion circuit  101  is a half-bridge high-frequency inversion circuit, M SP =0.5. If the high-frequency inversion circuit  101  is a full bridge high frequency inversion circuit, M SP =1. A value of M SR  varies with a working mode of the hybrid rectification circuit  103 . For example, if the hybrid rectification circuit  103  works in the full-bridge rectification mode, M SR =1. If the hybrid rectification circuit  103  works in the voltage doubling rectification mode, M SR =2 The resonant converter may adjust a frequency of a switching transistor to change a value of M LLC , so that V o  in case of different V in  can be obtained. 
       FIG.  11    is a diagram of a working principle of a resonant converter according to an embodiment. As shown in  FIG.  11   , different from a conventional soft switch technology, the LLC resonant converter may adjust the output voltage of the resonant converter by changing a working frequency of a pulse input into the LLC resonant tank network through pulse frequency modulation (PFM). V in  and the high-frequency inversion circuit  101  may be combined to be jointly simplified as an excitation source U i  of the LLC resonant tank network in  FIG.  11   . The voltage conversion circuit  104 , the hybrid rectification circuit  103 , the load capacitor, and the load resistor may be combined to be equivalent to an equivalent resistor R 1  in  FIG.  11   . The LLC resonant tank network  102  is equivalent to an LLC resonant tank network in  FIG.  11   . 
     The equivalent circuit may be deduced and calculated by using a fundamental component analysis (FHA) method. It may be understood from a working feature of the resonant circuit that a series resonance frequency of a resonant inductor L 1  and a resonant capacitor C 1  is 
     
       
         
           
             
               
                 f 
                 r 
               
               = 
               
                 1 
                 
                   2 
                   ⁢ 
                   π 
                   ⁢ 
                   
                     
                       
                         L 
                         1 
                       
                       ⁢ 
                       
                         C 
                         1 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
     and an output equivalent resistance of the LLC resonant tank network is obtained as 
     
       
         
           
             
               R 
               ac 
             
             = 
             
               
                 
                   π 
                   2 
                 
                 8 
               
               ⁢ 
               
                 R 
                 1 
               
             
           
         
       
     
     according to Ohm&#39;s law. 
     When the hybrid rectification circuit  103  works in the full-bridge rectification mode, the equivalent resistance is 
     
       
         
           
             
               R 
               
                 1 
                 ⁢ 
                 
                   _ 
                   ⁢ 
                   full 
                 
               
             
             = 
             
               
                 
                   V 
                   o 
                   2 
                 
                 
                   P 
                   o 
                 
               
               . 
             
           
         
       
     
     An expression of M LLC  is as follows: 
     
       
         
           
             
               M 
               
                 LLC 
                 ⁢ 
                 _ 
                 ⁢ 
                 full 
               
             
             = 
             
               1 
               
                 
                   
                     
                       [ 
                       
                         1 
                         + 
                         λ 
                         - 
                         
                           λ 
                           
                             
                               ( 
                               
                                 
                                   f 
                                   s 
                                 
                                 
                                   f 
                                   r 
                                 
                               
                               ) 
                             
                             2 
                           
                         
                       
                       ] 
                     
                     2 
                   
                   + 
                   
                     
                       
                         Q 
                         full 
                         2 
                       
                       ( 
                       
                         
                           
                             f 
                             s 
                           
                           
                             f 
                             r 
                           
                         
                         - 
                         
                           
                             f 
                             r 
                           
                           
                             f 
                             s 
                           
                         
                       
                       ) 
                     
                     2 
                   
                 
               
             
           
         
       
       
         
           
             
               
                 where 
                 ⁢ 
                 
                     
                      
                 
                 ⁢ 
                 
                   f 
                   r 
                 
               
               = 
               
                 1 
                 
                   2 
                   ⁢ 
                   π 
                   ⁢ 
                   
                     
                       
                         L 
                         1 
                       
                       ⁢ 
                       
                         C 
                         1 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
     a quality factor is 
     
       
         
           
             
               
                 Q 
                 full 
               
               = 
               
                 
                   Z 
                   1 
                 
                 
                   R 
                   ac 
                 
               
             
               
             , 
             
               
                 and 
                 ⁢ 
                     
                 
                   Z 
                   1 
                 
               
               = 
               
                 
                   
                     
                       L 
                       1 
                     
                     
                       C 
                       1 
                     
                   
                 
                 . 
               
             
           
         
       
     
     When the hybrid rectification circuit  103  works in the voltage doubling rectification mode, the equivalent resistance is 
     
       
         
           
             
               R 
               
                 1 
                 ⁢ 
                 
                   _ 
                   ⁢ 
                   double 
                 
               
             
             = 
             
               
                 
                   ( 
                   
                     
                       V 
                       o 
                     
                     2 
                   
                   ) 
                 
                 
                   P 
                   o 
                 
               
               = 
               
                 
                   
                     R 
                     
                       1 
                       ⁢ 
                       
                         _ 
                         ⁢ 
                         full 
                       
                     
                   
                   4 
                 
                 . 
               
             
           
         
       
     
     An expression of M LLC  is as follows: 
     
       
         
           
             
               M 
               
                 LLC 
                 ⁢ 
                 _ 
                 ⁢ 
                 double 
               
             
             = 
             
               1 
               
                 
                   
                     
                       [ 
                       
                         1 
                         + 
                         λ 
                         - 
                         
                           λ 
                           
                             
                               ( 
                               
                                 
                                   f 
                                   s 
                                 
                                 
                                   f 
                                   r 
                                 
                               
                               ) 
                             
                             2 
                           
                         
                       
                       ] 
                     
                     2 
                   
                   + 
                   
                     
                       
                         ( 
                         
                           Q 
                           double 
                         
                         ) 
                       
                       2 
                     
                     ⁢ 
                     
                       
                         ( 
                         
                           
                             
                               f 
                               s 
                             
                             
                               f 
                               r 
                             
                           
                           - 
                           
                             
                               f 
                               r 
                             
                             
                               f 
                               s 
                             
                           
                         
                         ) 
                       
                       2 
                     
                   
                 
               
             
           
         
       
       
         
           
             
               
                 where 
                 ⁢ 
                 
                     
                      
                 
                 ⁢ 
                 λ 
               
               = 
               
                 
                   L 
                   1 
                 
                 
                   L 
                   2 
                 
               
             
             , 
             
               
                 f 
                 r 
               
               = 
               
                 1 
                 
                   2 
                   ⁢ 
                   π 
                   ⁢ 
                   
                     
                       
                         L 
                         1 
                       
                       ⁢ 
                       
                         C 
                         1 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
     a quality factor is 
     
       
         
           
             
               
                 Q 
                 double 
               
               = 
               
                 4 
                 ⁢ 
                 
                   Q 
                   full 
                 
               
             
             , 
             
               
                 and 
                 ⁢ 
                     
                 
                   Z 
                   1 
                 
               
               = 
               
                 
                   
                     
                       L 
                       1 
                     
                     
                       C 
                       1 
                     
                   
                 
                 . 
               
             
           
         
       
     
     In an embodiment, a gain range of the resonant converter may be adjusted in a range of 1-2. An example in which the high-frequency inversion circuit  101  is a full-bridge high-frequency inversion circuit and the hybrid rectification circuit  103  is a circuit that includes a second capacitor and a rectifier bridge that includes four switching transistors is used for detailed description. 
       FIG.  12    is a schematic diagram of another gain curve according to an embodiment. As shown in  FIG.  12   , in the figure, a horizontal coordinate is a normalized frequency f s /f r  of a switching transistor, and a vertical coordinate is a gain of the resonant converter, where 
     
       
         
           
             
               f 
               min 
             
             = 
             
               
                 f 
                 
                   s 
                   ⁢ 
                   _ 
                   ⁢ 
                   min 
                 
               
               
                 f 
                 
                   r 
                   ⁢ 
                   _ 
                   ⁢ 
                   min 
                 
               
             
           
         
       
     
     represents a minimum normalized frequency of the switching transistor, and 
     
       
         
           
             
               f 
               max 
             
             = 
             
               
                 f 
                 
                   s 
                   ⁢ 
                   _ 
                   ⁢ 
                   max 
                 
               
               
                 f 
                 
                   r 
                   ⁢ 
                   _ 
                   ⁢ 
                   max 
                 
               
             
           
         
       
     
     represents a maximum normalized frequency of the switching transistor. A curve  1  is an adjustable gain curve existing when the high-frequency inversion circuit  101  works in a full-bridge high-frequency inversion mode and the hybrid rectification circuit  103  works in the full-bridge rectification mode. A curve  2  is an adjustable gain curve existing when the high-frequency inversion circuit  101  works in a full-bridge high-frequency inversion mode and the hybrid rectification circuit  103  works in the voltage doubling rectification mode. It may be understood from  FIG.  12    that, when the hybrid rectification circuit  103  works in the full-bridge rectification mode, the gain of the resonant converter changes between M min , and M middle ; and when the hybrid rectification circuit  103  works in the voltage doubling rectification mode, the gain of the resonant converter changes between M middle  and M max . 
     In an embodiment, when an adjustment range of the output voltage of the resonant converter is continuous, that is, when the gain range of the resonant converter changes in a range of M min -M middle -M max , M middle  is selected in a common adjustable area between the curve  1  and the curve  2 . Shapes and relative locations of the curve  1  and the curve  2  in  FIG.  12    can be adjusted by designing a circuit parameter. 
     In another embodiment,  FIG.  13    is a schematic diagram of still another gain curve according to an embodiment. As shown in  FIG.  13   , when an adjustment range of the output voltage of the resonant converter is segmented, that is, when the gain range of the resonant converter changes in a range of M min -M middle1  and in a range of M middle2 -M max , where M middle1  and M middle2  are different critical-point gains, M middle1  and M middle2  are selected in different adjustable areas of the curve  1  and the curve  2 . Shapes and relative locations of the curve  1  and the curve  2  in  FIG.  13    can be adjusted by designing a circuit parameter. 
     In this embodiment, when the direct current voltage adjustment gain falls within the first threshold range (M min -M middle  in  FIG.  12    or M min -M middle1  in  FIG.  13   ), the hybrid rectification circuit  103  may work in the full-bridge rectification mode to convert the second alternating current voltage into the second direct current voltage for output. When the direct current voltage adjustment gain falls within the second threshold range (M middle -M max  in  FIG.  12    or M middle2 -M max  in  FIG.  13   ), the hybrid rectification circuit  103  may work in the voltage doubling rectification mode to convert the second alternating current voltage into the second direct current voltage for output. 
     When the direct current voltage adjustment gain falls within the first threshold range, the gain of the high-frequency inversion circuit  101  is M SP =1, the gain M LLC  of the LLC resonant tank network  102  may vary with the frequency f of the switching transistor, and the gain of the hybrid rectification circuit  103  is M SR =1.  FIG.  14 A  is a schematic diagram of a waveform according to an embodiment. As shown in  FIG.  14 A , in this working state, drive signals of S p1 , S R1 , S p4 , and S R4  are in a same phase, drive signals of S p2 , S R2 , S p3 , and S R4  are in a same phase, and phases of drive signals of S p , and S p2  are complementary to each other. 
     When the direct current voltage adjustment gain falls within the second threshold range, one of the switching transistors of the rectifier bridge is always turned off, and another switching transistor of a same bridge arm is always turned on, so that the hybrid rectification circuit  103  can be switched from the full-bridge rectification mode to the voltage doubling rectification mode for working. 
     In a first implementation,  FIG.  14 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  14 B , S R1  is always turned off, and S R2  is always turned on, so that the schematic diagram of the structure of the resonant converter shown in  FIG.  14 C  can be obtained. In this case, the hybrid rectification circuit  103  works in the voltage doubling rectification mode, and the gain thereof is M SR =2.  FIG.  14 D  is a schematic diagram of another waveform according to an embodiment. As shown in  FIG.  14 D , in this working state, drive signals of S p1 , S p4 , and S R4  are in a same phase, drive signals of S p2 , S p3 , and S R3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
     In a second implementation,  FIG.  14 E  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  14 E , S R2  is always turned off, and S R1  is always turned on, so that the schematic diagram of the structure of the resonant converter shown in  FIG.  14 F  can be obtained. In this case, the hybrid rectification circuit  103  works in the voltage doubling rectification mode, and the gain thereof is M SR =2.  FIG.  14 G  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  14 G , in this working state, drive signals of S p1 , S p4 , and S R4  are in a same phase, drive signals of S p2 , S p3 , and S R3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
     In implementation, the high-frequency inversion circuit  101  of the resonant converter may be a full-bridge high-frequency inversion circuit. When the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit  103  may work in the full-bridge rectification mode. When the direct current voltage adjustment gain falls within the second threshold range, one of the switching transistors of the rectifier bridge is always turned off, and another switching transistor of a same bridge arm is always turned on, so that the hybrid rectification circuit  103  can be switched from the full-bridge rectification mode to the voltage doubling rectification mode for working. The hybrid rectification circuit  103  is switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     In another implementation,  FIG.  15 A  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  15 A , the rectifier bridge may include a first diode D R1 , a second diode D R2 , a third diode D R3 , a fourth diode D R4 , and one switching transistor S R . A negative electrode of D R1  is coupled to a negative electrode of D R3 . A positive electrode of D R1  is separately coupled to the second output end of the LLC resonant tank network  102 , a negative electrode of D R2 , and a drain of S R . A positive electrode of D R2  is separately coupled to a source of S R  and a positive electrode of D R4 . A positive electrode of D R3  is separately coupled to the second end of C p  and a negative electrode of D R4 . The negative electrode of D R3  and the positive electrode of D R4  are used as the output end of the resonant converter. 
     In an embodiment, a gain range of the resonant converter may be adjusted in a range of 1-2. An example in which the high-frequency inversion circuit is a full-bridge high-frequency inversion circuit and the rectifier bridge of the hybrid rectification circuit is a circuit that includes four diodes and one switching transistor is used for detailed description. 
     In this embodiment, when the direct current voltage adjustment gain falls within the first threshold range (M min -M middle  in  FIG.  12    or M min -M middle1  in  FIG.  13   ), the hybrid rectification circuit  103  may work in the full-bridge rectification mode to convert the second alternating current voltage into the second direct current voltage for output. When the direct current voltage adjustment gain falls within the second threshold range (M middle -M max  in  FIG.  12    or M middle2 -M max  in  FIG.  13   ), the hybrid rectification circuit  103  may work in the voltage doubling rectification mode to convert the second alternating current voltage into the second direct current voltage for output. 
     When the direct current voltage adjustment gain falls within the first threshold range,  FIG.  15 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  15 B , the switching transistor S R  is always turned off, so that the schematic diagram of the structure of the resonant converter shown in  FIG.  15 C  can be obtained. In this case, the gain of the high-frequency inversion circuit  101  is M SR =1, the gain M LLC  of the LLC resonant tank network  102  may vary with the frequency f of the switching transistor, and the gain of the hybrid rectification circuit  103  is M SR =1.  FIG.  15 D  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  15 D , in this working state, drive signals of S p1  and S p4  are in a same phase, drive signals of S p2  and S p3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
     When the direct current voltage adjustment gain falls within the second threshold range, one of the diodes of the rectifier bridge is always turned off, and the switching transistor S R  is always turned on, so that the hybrid rectification circuit  103  can be switched from the full-bridge rectification mode to the voltage doubling rectification mode for working. 
       FIG.  15 E  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  15 E , D R1  is always turned off, and S R  is always turned on, so that the schematic diagram of the structure of the resonant converter shown in  FIG.  15 F  can be obtained. In this case, the hybrid rectification circuit  103  works in the voltage doubling rectification mode, and the gain thereof is M SR =2.  FIG.  15 G  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  15 G , in this working state, drive signals of S p1  and S p4  are in a same phase, drive signals of S p2  and S p3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
     In implementation, the high-frequency inversion circuit  101  of the resonant converter may be a full-bridge high-frequency inversion circuit. When the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit  103  works in the full-bridge rectification mode. When the direct current voltage adjustment gain falls within the second threshold range, one of the diodes of the rectifier bridge is always turned off, and the switching transistor is always turned on, so that the hybrid rectification circuit  103  can be switched from the full-bridge rectification mode to the voltage doubling rectification mode for working. The hybrid rectification circuit  103  is switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     In an embodiment, a gain range of the resonant converter may be adjusted in a range of 0.5-2. The following is described in detail. 
       FIG.  16 A  is a schematic diagram of still another gain curve according to an embodiment. As shown in  FIG.  16 A , a curve  1  is a gain curve existing when the high-frequency inversion circuit  101  works in a half-bridge inversion mode and the hybrid rectification circuit  103  works in the full-bridge rectification mode. A curve  2  is a gain curve existing when the high-frequency inversion circuit  101  works in a half-bridge inversion mode and the hybrid rectification circuit  103  works in the voltage doubling rectification mode. A curve  3  is a gain curve existing when the high-frequency inversion circuit  101  works in a full-bridge inversion mode and the hybrid rectification circuit  103  works in the full-bridge rectification mode. A curve  4  is a gain curve existing when the high-frequency inversion circuit  101  works in a full-bridge inversion mode and the hybrid rectification circuit  103  works in the voltage doubling rectification mode. 
     In this embodiment,  FIG.  16 B  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  16 B , the high-frequency inversion circuit  101  is a half-bridge high-frequency inversion circuit, and a gain thereof is M SP =0.5. The gain M LLC  of the LLC resonant tank network  102  varies with the frequency f s  of the switching transistor. When the direct current voltage adjustment gain falls within the first threshold range (M min -M middle1  in  FIG.  16 A ), the hybrid rectification circuit  103  may work in the full-bridge rectification mode, and the gain thereof is M SR =1.  FIG.  16 C  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  16 C , in this working state, phases of drive signals of S p3  and S p4  are complementary to each other, and drive signals of S p1  and S R  are in a same phase. 
       FIG.  16 D  is a schematic diagram of a structure of still another resonant converter according to an embodiment. As shown in  FIG.  16 D , when the direct current voltage adjustment gain falls within the second threshold range (M middle1 -M middle2  in  FIG.  16 A ), the hybrid rectification circuit  103  may work in the voltage doubling rectification mode, and the gain thereof is M SR=2    FIG.  16 E  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  16 E , in this working state, phases of drive signals of S p3  and S p4  are complementary to each other, and drive signals of S p2  and S R  are in a same phase. 
     As shown in  FIG.  15 C , the high-frequency inversion circuit  101  is a full-bridge high-frequency inversion circuit, and a gain thereof is M SP , =1. The gain M LLC  of the LLC resonant tank network  102  varies with the frequency f s  of the switching transistor. When the direct current voltage adjustment gain falls within the first threshold range (M middle2 -M middle3  in  FIG.  16 A ), the hybrid rectification circuit  103  may work in the full-bridge rectification mode, and the gain thereof is M SR =1.  FIG.  16 F  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  16 F , in this working state, drive signals of S p1  and S p4  are in a same phase, drive signals of S p2  and S p3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
     As shown in  FIG.  15 F , when the direct current voltage adjustment gain falls within the second threshold range (M middle3 -M max  in  FIG.  16 A ), the hybrid rectification circuit  103  may work in the voltage doubling rectification mode, and the gain thereof is M SR =2.  FIG.  16 G  is a schematic diagram of still another waveform according to an embodiment. As shown in  FIG.  16 G , in this working state, drive signals of S p , and S p4  are in a same phase, drive signals of S p2  and S p3  are in a same phase, and phases of drive signals of S p1  and S p2  are complementary to each other. 
       FIG.  17    is a schematic diagram of a structure of still another resonant converter according to an embodiment.  FIG.  17    is obtained by optimizing the resonant converter shown in  FIG.  6   . 
     In another embodiment, the voltage conversion circuit  104  is shown in  FIG.  17   . Two ends of a primary side of T are respectively coupled to two output ends of the LLC resonant tank network  102 , two ends of L m  are respectively connected to two ends of a secondary side of T, and the two ends of L m  are respectively coupled to two input ends of the hybrid rectification circuit  103 . 
     In another embodiment,  FIG.  18    is a schematic diagram of a structure of coupling an LLC resonant tank network to a voltage conversion circuit according to an embodiment. As shown in  FIG.  18   , in an implementation, a first end of L r1  is coupled to a first output end of the high-frequency inversion circuit  101 . A first end of C r  is coupled to a second output end of the high-frequency inversion circuit  101 . A second end of L r1  is coupled to a first end of L r2 . A second end of C r  is coupled to a second end of L r2 . The first end of L r2  is coupled to a first input end of the voltage conversion circuit  104 . The second end of L r2  is coupled to a second input end of the voltage conversion circuit  104 . In another implementation, a first end of L r1  is coupled to a first output end of the high-frequency inversion circuit  101 . A second end of L r1  is coupled to a first input end of the voltage conversion circuit  104 . A second output end of the high-frequency inversion circuit  101  is coupled to a second input end of the voltage conversion circuit  104 . A first end of L r2  is separately coupled to a first output end of the voltage conversion circuit  104  and a first input end of the hybrid rectification circuit  103 . A first end of C r  is coupled to a second output end of the voltage conversion circuit  104 . The second end of L r2  is separately coupled to a second end of C r  and a second input end of the hybrid rectification circuit  103 . In still another implementation, a first input end of the voltage conversion circuit  104  is coupled to a first output end of the inversion circuit  101 . A first end of C r  is coupled to a second output end of the high-frequency inversion circuit  101 . A second end of C r  is coupled to a second input end of the voltage conversion circuit  104 . A first end of L r1  is coupled to a first output end of the voltage conversion circuit  104 . A second end of L r1  is separately coupled to a first end of L r2  and a first input end of the hybrid rectification circuit  103 . A second end of L r2  is separately coupled to a second output end of the voltage conversion circuit  104  and a second input end of the hybrid rectification circuit  103 . 
     In implementation, the hybrid rectification circuit  103  of the resonant converter may be switched between the full-bridge rectification working mode and the voltage doubling rectification working mode, so that the resonant converter has a wider gain range in a same switching frequency range, to reduce a reactive power cycle loss, so as to implement high conversion efficiency during wide input-output voltage power conversion. 
     It should be noted that the switching transistor provided in this embodiment may be an insulated gate bipolar transistor (IGBT), a metal-oxide semiconductor field-effect transistor (MOSFET, a transistor, or the like. This is not limited. 
       FIG.  19    is a schematic flowchart of a voltage conversion method according to an embodiment. The voltage conversion method is applied to a resonant converter. As shown in  FIG.  4   , the resonant converter may include a high-frequency inversion circuit, an LLC resonant tank network, and a hybrid rectification circuit. As shown in  FIG.  19   , the voltage conversion method may include the following steps: 
       1901 . Convert a first direct current voltage into a first alternating current voltage by using the high-frequency inversion circuit. 
       1902 . Adjust the first alternating current voltage by using the LLC resonant tank network to obtain a second alternating current voltage. 
       1903 . Enable, when a direct current voltage adjustment gain falls within a first threshold range, the hybrid rectification circuit to work in a full-bridge rectification mode to convert the second alternating current voltage into a second direct current voltage for output; or enable, when a direct current voltage adjustment gain falls within a second threshold range, a hybrid rectification circuit to work in a voltage doubling rectification mode to convert the second alternating current voltage into a second direct current voltage for output. 
     The resonant converter may further include a voltage conversion circuit. In an embodiment, the method may further include: adjusting the first alternating current voltage by using the voltage conversion circuit to obtain a third alternating current voltage. The adjusting the first alternating current voltage by using the LLC resonant tank network to obtain a second alternating current voltage includes: adjusting the third alternating current voltage by using the LLC resonant tank network to obtain the second alternating current voltage. The method corresponds to the resonant converter shown in  FIG.  5   . 
     In another embodiment, the method may further include: adjusting the second alternating current voltage by using the voltage conversion circuit to obtain a fourth alternating current voltage. The enabling, when a direct current voltage adjustment gain falls within a first threshold range, the hybrid rectification circuit to work in a full-bridge rectification mode to convert the second alternating current voltage into a second direct current voltage for output includes: 
     enabling, when the direct current voltage adjustment gain falls within the first threshold range, the hybrid rectification circuit to work in the full-bridge rectification mode to convert the fourth alternating current voltage into the second direct current voltage. 
     The enabling, when a direct current voltage adjustment gain falls within a second threshold range, the hybrid rectification circuit to work in a voltage doubling rectification mode to convert the second alternating current voltage into a second direct current voltage for output includes: 
     enabling, when the direct current voltage adjustment gain falls within the second threshold range, the hybrid rectification circuit to work in the voltage doubling rectification mode to convert the fourth alternating current voltage into the second direct current voltage for output. The method corresponds to the resonant converter shown in  FIG.  6   . 
     For detailed descriptions of each step in the foregoing method, refer to the foregoing related descriptions. Details are not described herein again. 
     The objectives, solutions, and benefits are further described in detail in the foregoing embodiments. It should be understood that the foregoing descriptions are merely implementations, but are not intended to limit the scope of the embodiments. Any modification, equivalent replacement, or improvement shall fall within the scope of the embodiments.