Patent Publication Number: US-7907736-B2

Title: Acoustic correction apparatus

Description:
This application is a continuation of U.S. patent application Ser. No. 09/411,143, filed on Oct. 4, 1999, now U.S. Pat. No. 7,031,474, the entirety of which is hereby incorporated herein by reference. 
    
    
     FIELD OF THE INVENTION 
     This invention relates generally to audio enhancement systems, and especially those systems and methods designed to improve the realism of stereo sound reproduction. More particularly, this invention relates to an apparatus for overcoming the acoustic imaging and frequency response deficiencies of a sound system as perceived by a listener. 
     BACKGROUND OF THE INVENTION 
     In a sound reproduction environment, various factors may serve to degrade the quality of reproduced sound as perceived by a listener. Such factors distinguish the sound reproduction from that of an original sound stage. One such factor is the location of loudspeakers in a sound stage, which, if inappropriately placed, may lead to a distorted sound-pressure response over the audible frequency spectrum. The placement of loudspeakers also affects the perceived width of a soundstage. For example, loudspeakers act as point sources of sound limiting their ability to reproduce reverberant sounds that are easily perceived in a live sound stage. In fact, the perceived sound stage width of many audio reproduction systems is limited to the distance separating a pair of loudspeakers when placed in front of a listener. Another factor degrading the quality of reproduced sound may result from microphones, which record sound differently from the way the human hearing system perceives sound. In an attempt to overcome the factors, which degrade the quality of reproduced sound, countless efforts have been expended to alter the characteristics of a sound reproduction environment to mimic that heard by a listener in a live sound stage. 
     Some efforts at stereo image enhancement have focused on the acoustic abilities and limitations of the human ear. The human ear&#39;s auditory response is sensitive to sound intensity, phase differences between certain sounds, the frequency of the sound itself, and the direction from which sound emanates. Despite the complexity of the human auditory system, the frequency response of the human ear is relatively constant from person to person. 
     When sound waves having a constant sound pressure level across all frequencies are directed at a listener from a single location, the human ear will react differently to the individual frequency components of the sound. For example, when sound of equal sound pressure is directed towards a listener from in front of the listener, the pressure level created within the listener&#39;s ear by a sound of 1000 hertz will be different from that of 2000 hertz. 
     In addition to frequency sensitivity, the human auditory system reacts differently to sounds impinging upon the ear from various angles. Specifically, the sound pressure level within the human ear will vary with the direction of sound. The shape of the outer ear, or pinna, and the inner ear canal are largely responsible for the frequency contouring of sounds as a function of direction. 
     The human auditory response is sensitive to both azimuth and elevation changes of a sound&#39;s origin. This is particularly true for complex sound signals, i.e., those having multiple frequency components, and for higher frequency components in general. The variance in sound pressure among the frequency components within the ear is interpreted by the brain to provide indications of a sound&#39;s origin. When a recorded sound is reproduced, the directional cues to the sound&#39;s origin, as interpreted by the ear from sound pressure information, will thus be dependent upon the actual location of loudspeakers that reproduce the sound. 
     A constant sound pressure level, i.e., a “flat” sound pressure versus frequency response, can be obtained at the ears of a listener from loudspeakers positioned directly in front of the listener. Such a response is often desirable to achieve a realistic sound image. However, the quality of a set of loudspeakers may be less than ideal, and they may not be placed in the most acoustically-desirable location. Both such factors often lead to disrupted sound pressure characteristics. Sound systems of the prior art have disclosed methods to “correct” the sound pressure emanating from loudspeakers to create a spatially correct response thereby improving the resulting sound image. 
     To achieve a more spatially correct response for a given sound system, it is known to select and apply head-related-transfer-functions (HRTFs) to an audio signal. HRTFs are based on the acoustics of the human hearing system. Application of an HRTF is used to adjust the amplitudes of portions of the audio signal to compensate for spatial distortion. HRTF-based principles may also be used to relocate a stereo image from non-optimally placed loudspeakers. 
     A second type of deficiency often occurs because it is difficult to adequately reproduce low-frequency sounds such as bass. Various conventional approaches to improving the output of low-frequency sounds include the use of higher quality loudspeakers with greater cone areas, larger magnets, larger housings, or greater cone excursion capabilities. In addition, conventional systems have attempted to reproduce low-frequency sounds with resonant chambers and horns that match the acoustic impedance of the loudspeaker to the acoustic impedance of free space surrounding the loudspeaker. 
     Not all systems, however, can simply use more expensive or more powerful loudspeakers to reproduce low-frequency sounds. For example, some conventional sound systems such as compact audio systems and multimedia computer systems rely on small loudspeakers. In addition, to conserve costs, many audio systems use less accurate loudspeakers. Such loudspeakers typically do not have the capability to properly reproduce low-frequency sounds and consequently, the sounds are typically not as robust or enjoyable as systems that more accurately reproduce low-frequency sounds. 
     Some conventional enhancement systems attempt to compensate for poor reproduction of low-frequency sounds by amplifying the low-frequency signals prior to inputting the signals into the loudspeakers. Amplifying the low-frequency signals delivers a greater amount of energy to the loudspeakers, which in turn, drives the loudspeakers with greater forces. Such attempts to amplify the low-frequency signals, however, can result in overdriving the loudspeakers. Unfortunately, overdriving the loudspeakers can increase the background noise, introduce distracting distortions, and damage the loudspeakers. 
     Still other conventional systems, in an attempt to compensate for the lack of the lower-frequencies, distort the reproduction of the higher frequencies in ways that add undesirable sound coloration. 
     A third difficulty arises because sounds emanating from multiple locations are often not properly reproduced in an audio system. One approach directed to improving the reproduction of sound includes surround sound systems that have multiple recording tracks. The multiple recording tracks are used to record the spatial information associated with sounds that emanate from multiple locations. 
     For example, in a surround sound system, some of the recording tracks contain sounds that originate from in front of the listener, while other recording tracks contain sounds, which originate from behind the listener. When multiple loudspeakers are placed around the listener, the audio information contained in the recording tracks makes the produced sounds appear more realistic to the listener. Such systems, however, are typically more expensive than systems, which do not use multiple recording tracks and multiple speaker arrangements. 
     To conserve costs, many conventional two-speaker systems attempt to simulate a surround sound experience by introducing unnatural time-delays or phase-shifts between left and right signal sources. Unfortunately, such systems often suffer from unrealistic effects in the reproduced sound. 
     Other known sound enhancement techniques operate on what are called “sum” and “difference” signals. The sum signal, which is also called the monophonic signal, is the sum of the left and right signals. This can be conceptualized as adding or combining the left and right signals (L+R). 
     The difference signal, on the other hand, represents the difference between the two left and right audio signals. This is best conceptualized as subtracting the right signal from the left signal (L−R). The difference signal is also often called the ambient signal. 
     It is known that modifying certain frequencies in the difference signal can widen the perceived sound projected from the left and right loudspeakers. The widened sound image typically results from altering the reverberant sounds, which are present in the difference signal. 
     The circuitry that generates the sum and difference signals, however, generates the sum and difference signals by processing of the left and right input signals. Furthermore, once the circuitry generates the sum and difference signals, additional circuitry then separately processes and recombines the sum and difference signals in order to produce an enhanced sound effect. 
     Typically, the creation and processing of the sum and difference signal are accomplished with digital signal processors, operational amplifiers and the like. Such implementations usually require complicated circuitry that increases the cost of such systems. Thus, despite the contributions from the prior art, there exists a need for a simplified audio enhancement system that reduces costs associated with producing an enhanced listening experience. 
     SUMMARY OF THE INVENTION 
     The present invention solves these and other problems by providing a signal processing technique that significantly improves the image size, bass performance and dynamics of an audio system, surrounding the listener with an engaging and powerful representation of the audio performance. It improves the listening experience for a variety of applications, including computer, multimedia, televisions, boom-boxes, automobiles, home audio, and portable audio systems. In one embodiment, the sound correction system corrects for the apparent placement of the loudspeakers, the image created by the loudspeakers, and the low frequency response produced by the loudspeakers. In one embodiment, the sound correction system enhances spatial and frequency response characteristics of sound reproduced by two or more loudspeakers. The audio correction system includes an image correction module that corrects the listener-perceived vertical image of the sound reproduced by the loudspeakers, a bass enhancement module that improves the listener-perceived bass response of the loudspeakers, and an image enhancement module that enhances the listener-perceived horizontal image of the apparent sound stage. 
     In one embodiment, three processing techniques are used. Spatial cues responsible for positioning sound outside the boundaries of the speaker are equalized using Head Related Transfer Functions (HRTFs). These HRTF correction curves account for how the brain perceives the location of sounds to the sides of a listener even when played back through speakers in front of the listener. As a result, the presentation of instruments and vocalists occur in their proper place, with the addition of indirect and reflected sounds all about the room. A second set of HRTF correction curves expands and elevates the apparent size of the stereo image, such that the sound stage takes on a scale of immense proportion compared to the speaker locations. Finally, bass performance is enhanced through a psychoacoustic technique that restores the perception of low frequency fundamental tones by dynamically augmenting harmonics that the speaker can more easily reproduce. 
     The acoustic correction system, and the associated methods of operation, provide a sophisticated and effective system for improving the vertical, horizontal, and spectral sound image in an imperfect reproduction environment. In one embodiment, the system first corrects the vertical image produced by the loudspeakers, then the bass is enhanced, and finally, the horizontal image is corrected. The vertical image enhancement typically includes some emphasis of the lower frequency portions of the sound, and thus providing vertical enhancement before bass enhancement contributes to the overall effect of the bass enhancement processing. The bass enhancement provides some mixing of the common portions of the left and right portions of the low frequency information in a stereophonic signal (common-mode). By contrast, the horizontal image enhancement provides some enhancement and shaping of the differences between the left and right portions (differential-mode). Thus, in one embodiment, bass enhancement is advantageously provided before horizontal image enhancement in order to balance the common-mode and differential-mode portions of the stereophonic signal to produce a pleasing effect for the listener. 
     To achieve an improved stereo image in the vertical plane, an image correction device divides an input signal into first and second frequency ranges that collectively contain substantially all of the audio frequency spectrum. The frequency response characteristics of the input signal within the first and second frequency ranges are separately corrected and combined to create an output signal having a relatively flat frequency-response characteristic with respect to a listener. The level of frequency correction, i.e., sound-energy correction, is dependent upon the reproduction environment and tailored to overcome the acoustic limitations of such an environment. The design of the acoustic correction apparatus allows for easy and independent correction of the input signal within individual frequency ranges to achieve a spatially-corrected and relocated sound image. 
     Within an audio reproduction environment, loudspeakers may be poorly located, thereby adversely affecting a sound image perceived by the listener. For example, headphones often produce an unpleasing sound image because the transducers are located right next to the listener&#39;s ears. The acoustic correction apparatus of the present invention relocates the sound image to a more pleasing apparent position. 
     Through application of the acoustic correction apparatus, a stereo image generated from playback of an audio signal may be spatially corrected to convey a perceived source of origin having a vertical and/or horizontal position distinct from the position of the loudspeakers. The exact source of origin perceived by a listener will depend on the level of spatial correction. 
     Once a perceived sound origin is obtained through correction of spatial distortion, the corrected audio signal may be enhanced to provide an expanded stereo image. In accordance with one embodiment, stereo image enhancement of a relocated audio image takes into account acoustic principles of human hearing to envelop the listener in a realistic sound stage. In those sound reproduction environments where a listening position is relatively fixed, (such as the interior of an automobile, multimedia computer systems, bookshelf speaker systems, etc.) the amount of stereo image enhancement applied to the audio signal is partially determined by the actual position of the loudspeakers with respect to the listener. 
     In loudspeakers that do not reproduce certain low-frequency sounds, the invention creates the illusion that the missing low-frequency sounds do exist. Thus, a listener perceives low frequencies, which are below the frequencies the loudspeaker can actually accurately reproduce. This illusionary effect is accomplished by exploiting, in a unique manner, how the human auditory system processes sound. 
     One embodiment of the invention exploits how a listener mentally perceives music or other sounds. The process of sound reproduction does not stop at the acoustic energy produced by the loudspeaker, but includes the ears, auditory nerves, brain, and thought processes of the listener. Hearing begins with the action of the ear and the auditory nerve system. The human ear may be regarded as a delicate translating system that receives acoustical vibrations, converts these vibrations into nerve impulses, and ultimately into the “sensation” or perception of sound. 
     Advantageously, some embodiments of the invention exploit how the human ear processes overtones and harmonics of low-frequency sounds to create the perception that non-existent low-frequency sounds are being emitted from a loudspeaker. In some embodiments, the frequencies in higher-frequency bands are selectively processed to create the illusion of lower-frequency signals. In other embodiments, certain higher-frequency bands are modified with a plurality of filter functions. 
     In addition, some embodiments of the invention are designed to improve the low-frequency enhancement of popular audio program material, such as music. Most music is rich in harmonics. Accordingly, these embodiments can modify a wide variety of music types to exploit how the human ear processes low-frequency sounds. Advantageously, music in existing formats can be processed to produce the desired effects. 
     This new approach produces a number of significant advantages. Because a listener perceives low-frequency sounds, which do not actually exist, the need for large loudspeakers, greater cone excursions, or added horns is reduced. Thus, in one embodiment, small loudspeakers can appear as if they are emitting the low-frequency sounds of larger loudspeakers. As can be expected, this embodiment produces the perception of low-frequency audio such as bass, in sound environments that are too small for large loudspeakers. Large loudspeakers are benefited as well, by creating the perception that they are producing enhanced low-frequency sounds. 
     In addition, with one embodiment of the invention, the small loudspeakers in hand-held and portable sound systems can create a more enjoyable perception of low-frequency sounds. Thus, the listener need not sacrifice low-frequency sound quality for portability. 
     In one embodiment of the invention, lower-cost loudspeakers create the illusion of low-frequency sounds. Many low-cost loudspeakers cannot adequately reproduce low-frequency sounds. Rather than actually reproducing low-frequency sounds with expensive speaker housings, high performance components and large magnets, one embodiment uses higher frequency sounds to create the illusion of low-frequency sounds. As a result, lower-cost loudspeakers can be used to create a more realistic and robust listening experience. 
     Furthermore, in one embodiment, the illusion of low-frequency sounds creates a heightened listening experience that increases the realism of the sound. Thus, instead of the reproduction of the muddy or wobbly low-frequency sounds existing in many low-cost prior art systems, one embodiment of the invention reproduces sounds that are perceived to be more accurate and clear. Such low-cost audio and audio-visual devices can include, by way of example, radios, mobile audio systems, computer games, loudspeakers, compact disc (CD) players, digital versatile disc (DVD) players, multimedia presentation devices, computer sound cards, and the like. 
     In one embodiment, creating the illusion of low-frequency sounds requires less energy than actually reproducing the low-frequency sounds. Thus, systems, which operate on batteries, low-power environments, small speakers, multimedia speakers, headphones, and the like, can create the illusion of low-frequency sounds without consuming as much valuable energy as systems, which simply amplify or boost low-frequency sounds. 
     Other embodiments of the invention create the illusion of lower-frequency signals with specialized circuitry. These circuits are simpler than prior art low-frequency amplifiers and thus reduce the costs of manufacturing. Advantageously, these cost less than prior art sound enhancement devices that add complex circuitry. 
     Still other embodiments of the invention rely on a microprocessor, which implements the disclosed low-frequency enhancement techniques. In some cases, existing processing audio components can be reprogrammed to provide the disclosed unique low-frequency signal enhancement techniques of one or more embodiments of the invention. As a result, the costs of adding low-frequency enhancement to existing systems is significantly reduced. 
     In one embodiment, the sound enhancement apparatus receives one or more input signals, from a host system and in turn, generates one or more enhanced output signals. In particular, the two input signals are processed to provide a pair of spectrally enhanced output signals, that when played on a loudspeaker and heard by a listener, produce the sensation of extended bass. In one embodiment, the low-frequency audio information is modified in a different manner than the high-frequency audio information. 
     In one embodiment, the sound enhancement apparatus receives one or more input signals and generates one or more enhanced output signals. In particular, the input signals comprise waveforms having a first frequency range and a second frequency range. The input signals are processed to provide the enhanced output signals, that when played on a loudspeaker and heard by a listener, produce the sensation of extended bass. In addition, the embodiment may modify information in the first frequency range in a different manner than information in the second frequency range. In some embodiments, the first frequency range may be bass frequencies too low for the desired loudspeaker to reproduce and the second frequency range may be midbass frequencies that the loudspeaker can reproduce. 
     One embodiment modifies the audio information that is common to two stereo channels in a manner different from energy that is not common to the two channels. The audio information that is common to both input signals is referred to as the combined signal. In one embodiment, the enhancement system spectrally shapes the amplitude of the phase and frequencies in the combined signal in order to reduce the clipping that may result from high-amplitude input signals without removing the perception that the audio information is in stereo. 
     As discussed in more detail below, one embodiment of the sound enhancement system spectrally shapes the combined signal with a variety of filters to create an enhanced signal. By enhancing selected frequency bands within the combined signal, the embodiment provides a perceived loudspeaker bandwidth that is wider than the actual loudspeaker bandwidth. 
     One embodiment of the sound enhancement apparatus includes feedforward signal paths for the two stereo channels and three parallel filters for the combined signal path. Each of the four parallel filters comprises a sixth order bandpass filter consisting of three series connected biquad filters. The transfer functions for these four filters are specially selected to provide phase and/or amplitude shaping of various harmonics of the low-frequency content of an audio signal. The shaping unexpectedly increases the perceived bandwidth of the audio signal when played through loudspeakers. In another embodiment, the sixth order filters are replaced by lower order Chebychev filters. 
     Because the spectral shaping occurs on the combined signal, which is then combined with the stereo information in the feedforward paths, the frequencies in the combined signal can be altered such that both stereo channels are affected, and some signals in certain frequency ranges are coupled from one stereo channel to the other stereo channel. As a result, various embodiments create enhanced audio sound in an entirely unique, novel, and unexpected manner. 
     The sound enhancement apparatus may in turn, be connected to one or more subsequent signal processing stages. These subsequent stages may provide improved soundstage or spatial processing. The output signals can also be directed to other audio devices such as recording devices, power amplifiers, loudspeakers, and the like without affecting the operation of the sound enhancement apparatus. 
     The present invention also provides a unique differential perspective correction system to improve the horizontal aspects of the sound image. The differential perspective correction system enhances sound in an entirely different way than other sound enhancement devices. Advantageously, the perspective correction system embodiment can be used to enhance sound in a wide range of low-cost audio and audio-visual devices, which by way of example can include radios, mobile audio systems, computer games, multimedia presentation devices, and the like. 
     Broadly speaking, the differential perspective correction apparatus receives two input signals, from a host system and in turn, generates two enhanced output signals. In particular, the two input signals are processed collectively to provide a pair of spatially corrected output signals. In addition, one embodiment modifies the audio information that is common to both input signals in a different manner than the audio information, which is not common to both input signals. 
     Audio information that is common to both input signals is referred to as the common-mode information, or the common-mode signal. The common-mode audio information differs from a sum signal in that rather than containing the sum of the input signals, it contains only that audio information which exists in both input signals at any given instant in time. 
     In contrast, the audio information which is not common to both input signals is referred to as the differential information or the differential signal. Although the differential information is processed in a different manner than the common-mode information, the differential information is not a discrete signal. As discussed in more detail below, the differential perspective correction apparatus spectrally shapes the differential signal with a variety of filters to create an equalized differential signal. By equalizing selected frequency bands within the differential signal, the differential perspective correction apparatus widens a perceived sound image projected from a pair of loudspeakers placed in front of a listener. 
     Because the cross-over impedance networks equalize the frequency ranges in the differential input, the frequencies in the differential signal can be altered without affecting the frequencies in the common-mode signal. As a result, the audio sound is enhanced in an entirely unique and novel manner. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other aspects, features, and advantages of the present invention will be more apparent from the following particular description thereof presented in conjunction with the following drawings, wherein: 
         FIG. 1  is a block diagram of a stereo image correction system operatively connected to a stereo enhancement system and a bass enhancement system for creating a realistic stereo image from a pair of input stereo signals. 
         FIG. 2  is a diagram of a stereo system including a stereo receiver and two speakers. 
         FIG. 3  is a diagram of a typical multimedia computer system. 
         FIG. 4A  is a graphical representation of a desired sound-pressure versus frequency characteristic for an audio reproduction system. 
         FIG. 4B  is a graphical representation of a sound-pressure versus frequency characteristic corresponding to a first audio reproduction environment. 
         FIG. 4C  is a graphical representation of a sound-pressure versus frequency characteristic corresponding to a second audio reproduction environment. 
         FIG. 4D  is a graphical representation of a sound-pressure versus frequency characteristic corresponding to a third audio reproduction environment. 
         FIG. 5  is a schematic block diagram of an energy-correction system operatively connected to a stereo image enhancement system for creating a realistic stereo image from a pair of input stereo signals. 
         FIG. 6A  is a graphical representation of the various levels of signal modification provided by a low-frequency correction system in accordance with one embodiment. 
         FIG. 6B  is a graphical representation of the various levels of signal modification provided by a high-frequency correction system for boosting high-frequency components of an audio signal in accordance with one embodiment. 
         FIG. 6C  is a graphical representation of the various levels of signal modification provided by a high-frequency correction system for attenuating high-frequency components of an audio signal in accordance with one embodiment. 
         FIG. 6D  is a graphical representation of a composite energy-correction curve depicting the possible ranges of sound-pressure correction for relocating a stereo image. 
         FIG. 7  is a graphical representation of various levels of equalization applied to an audio difference signal to achieve varying amounts of stereo image enhancement. 
         FIG. 8A  is a diagram depicting the perceived and actual origins of sounds heard by a listener from loudspeakers placed at a first location. 
         FIG. 8B  is a diagram depicting the perceived and actual origins of sounds heard by a listener from loudspeakers placed at a second location. 
         FIG. 9  is a plot of the frequency response of a typical small loudspeaker system. 
         FIG. 10  illustrates the actual and perceived spectrum of a signal represented by two discrete frequencies. 
         FIG. 11  illustrates the actual and perceived spectrum of a signal represented by a continuous spectrum of frequencies. 
         FIG. 12A  illustrates a time waveform of a modulated carrier. 
         FIG. 12B  illustrates the time waveform of  FIG. 12A  after detection by a detector. 
         FIG. 13A  is a block diagram of a sound system with bass enhancement processing. 
         FIG. 13B  is a block diagram of a bass enhancement processor that combines multiple channels into a single bass channel. 
         FIG. 13C  is a block diagram of a bass enhancement processor that processes multiple channels separately. 
         FIG. 14  is a signal processing block diagram of a system that provides bass enhancement with selectable frequency response. 
         FIG. 15  is a plot of the transfer functions of the bandpass filters used in the signal processing diagram shown in  FIG. 14 . 
         FIG. 16  is a time-domain plot showing the time-amplitude response of the punch system. 
         FIG. 17  is a time-domain plot showing the signal and envelope portions of a typical bass note played by an instrument, wherein the envelope shows attack, decay, sustain and release portions. 
         FIG. 18  is a signal processing block diagram of a system that provides bass enhancement using a peak compressor and a bass punch system. 
         FIG. 19  is a time-domain plot showing the effect of the peak compressor on an envelope with a fast attack. 
         FIG. 20  is a conceptual block diagram of a stereo image (differential perspective) correction system. 
         FIG. 21  is a block diagram of a stereo image (differential perspective) correction system that does not develop explicit sum and difference signals. 
         FIG. 22  illustrates a graphical representation of the common-mode gain of the differential perspective correction system. 
         FIG. 23  is a graphical representation of the overall differential signal equalization curve of the differential perspective correction system. 
         FIG. 24  is a block diagram of one embodiment of a sound enhancement system that can be implemented on a single chip. 
         FIG. 25A  is a schematic diagram of a left channel of a vertical image enhancement block suitable for use in the system shown in  FIG. 24 . 
         FIG. 25B  is a schematic diagram of a right channel of a vertical image enhancement block suitable for use in the system shown in  FIG. 24 . 
         FIG. 26  is a schematic diagram of a bass enhancement block suitable for use in the system shown in  FIG. 24 . 
         FIG. 27  is a schematic diagram of a filter system suitable for use in the bass enhancement system shown in  FIG. 26 . 
         FIG. 28  is a schematic diagram of a compressor system suitable for use in the bass enhancement system shown in  FIG. 26 . 
         FIG. 29  is a schematic diagram of a horizontal image enhancement block suitable for use in the system shown in  FIG. 24 . 
         FIG. 30  is a schematic diagram of a differential perspective correction system that can be used as the stereo image enhancement system. 
         FIG. 31  shows a differential perspective correction system using one crossover network. 
         FIG. 32  is a schematic diagram of a differential perspective correction apparatus using two crossover networks. 
         FIG. 33  shows a differential perspective correction apparatus that allows a user to vary the amount of overall differential gain. 
         FIG. 34  illustrates a differential perspective correction apparatus that allows a user to vary the amount of common-mode gain. 
         FIG. 35  illustrates a differential perspective correction apparatus that has a first crossover network located between the emitters of the transistors of a differential pair and a second crossover network located between the collectors of the differential pair. 
         FIG. 36  shows a differential perspective correction apparatus with output buffers. 
         FIG. 37  shows a six opamp version of an image enhancement system. 
         FIG. 38  is a block diagram of a software embodiment of the acoustic correction system. 
         FIG. 39  is a plot of the transfer function of a 40 Hz bandpass filter for use with the block diagram shown in  FIG. 38 . 
         FIG. 40  is a plot of the transfer function of a 60 Hz bandpass filter for use with the block diagram shown in  FIG. 38 . 
         FIG. 41  is a plot of the transfer function of a 100 Hz bandpass filter for use with the block diagram shown in  FIG. 38 . 
         FIG. 42  is a plot of the transfer function of a 150 Hz bandpass filter for use with the block diagram shown in  FIG. 38 . 
         FIG. 43  is a plot of the transfer function of a 200 Hz bandpass filter for use with the block diagram shown in  FIG. 38 . 
         FIG. 44  is a plot of the transfer function of a lowpass filter for use with the block diagram shown in  FIG. 38 . 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  is a block diagram of an acoustic correction apparatus  120  comprising, in series, a stereo image correction system  122 , a bass enhancement system  101 , and a stereo image enhancement system  124 . The image correction system  122  provides a left stereo signal and a right stereo signal to the bass enhancement unit  101 . The bass enhancement unit outputs left and right stereo signals to respective left and right inputs of the stereo image enhancement device  124 . The stereo image enhancement system  124  processes the signals and provides a left output signal  130  and a right output signal  132 . The output signals  130  and  132  may in turn be connected to some other form of signal conditioning system, or they may be connected directly to loudspeakers or headphones (not shown). 
     When connected to loudspeakers, the correction system  120  corrects for deficiencies in the placement of the loudspeakers, the image created by the loudspeakers, and the low frequency response produced by the loudspeakers. The sound correction system  120  enhances spatial and frequency response characteristics of the sound reproduced by the loudspeakers. In the audio correction system  120 , the image correction module  122  corrects the listener-perceived vertical image of an apparent sound stage reproduced by the loudspeakers, the bass enhancement module  101  improves the listener-perceived bass response of the sound, and the image enhancement module  124  enhances the listener-perceived horizontal image of the apparent sound stage. 
     The correction apparatus  120  improves the sound reproduced by loudspeakers by compensating for deficiencies in the sound reproduction environment and deficiencies of the loudspeakers. The apparatus  120  improves reproduction of the original sound stage by compensating for the location of the loudspeakers in the reproduction environment. The sound-stage reproduction is improved in a way that enhances both the horizontal and vertical aspects of the apparent (i.e. reproduced) sound stage over the audible frequency spectrum. The apparatus  120  advantageously modifies the reverberant sounds that are easily perceived in a live sound stage such that the reverberant sounds are also perceived by the listener in the reproduction environment, even though the loudspeakers act as point sources with limited ability. The apparatus  120  also compensates for the fact that microphones often record sound differently from the way the human hearing system perceives sound. The apparatus  120  uses filters and transfer functions that mimic human hearing to correct the sounds produced by the microphone. 
     The sound system  120  adjusts the apparent azimuth and elevation point of a complex sound by using the characteristics of the human auditory response. The correction is used by the listener&#39;s brain to provide indications of the sound&#39;s origin. The correction apparatus  120  also corrects for loudspeakers that are placed at less than ideal conditions, such as loudspeakers that are not in the most acoustically-desirable location. 
     To achieve a more spatially correct response for a given sound system, the acoustic correction apparatus  120  uses certain aspects of the head-related-transfer-functions (HRTFs) in connection with frequency response shaping of the sound information to correct both the placement of the loudspeakers, to correct the apparent width and height of the sound stage, and to correct for inadequacies in the low-frequency response of the loudspeakers. 
     Thus, the acoustic correction apparatus  120  provides a more natural and realistic sound stage for the listener, even when the loudspeakers are placed at less than ideal locations and when the loudspeakers themselves are inadequate to properly reproduce the desired sounds. 
     The various sound corrections provided by the correction apparatus are provided in an order such that subsequent correction does not interfere with prior corrections. In one embodiment, the corrections are provided in a desirable order such that prior corrections provided by the apparatus  120  enhance and contribute to the subsequent corrections provided by the apparatus  120 . 
     In one embodiment, the correction apparatus  120  simulates a surround sound system with improved bass response. The correction apparatus  120  creates the illusion that multiple loudspeakers are placed around the listener, and that audio information contained in multiple recording tracks is provided to the multiple speaker arrangement. 
     The acoustic correction system  120  provides a sophisticated and effective system for improving the vertical, horizontal, and spectral sound image in an imperfect reproduction environment. The image correction system  122  first corrects the vertical image produced by the loudspeakers. Then the bass enhanced system  101  adjusts the low frequency components of the sound signal in a manner that enhances the low frequency output of small loudspeakers that do no provide adequate low frequency reproduction capabilities. Finally, the horizontal sound image is corrected by the image enhancement system  124 . 
     The vertical image enhancement provided by the image correction system  122  typically includes some emphasis of the lower frequency portions of the sound, and thus providing vertical enhancement before the bass enhancement system  101  contributes to the overall effect of the bass enhancement processing. The bass enhancement system  101  provides some mixing of the common portions of the left and right portions of the low frequency information in a stereophonic signal (common-mode). By contrast, the horizontal image enhancement provided by the image enhancement system  124  provides enhancement and shaping of the differences between the left and right portions (differential-mode) of the signal. Thus, in the correction system  120 , bass enhancement is advantageously provided before horizontal image enhancement in order to balance the common-mode and differential-mode portions of the stereophonic signal to produce a pleasing effect for the listener. 
     As disclosed above, the stereo image correction system  122 , the bass enhancement system  101 , and the stereo image enhancement system  124  cooperate to overcome acoustic deficiencies of a sound reproduction environment. The sound reproduction environments may be as large as a theater complex or as small as a portable electronic keyboard. The acoustic correction apparatus also provides major benefits for a multimedia computer systems (see e.g.,  FIG. 3 ), home audio, televisions, headphones, boom-boxes, automobiles, and the like. 
       FIG. 2  shows a stereophonic audio system having a receiver  220 . The receiver  220  provides a left channel signal to a left speaker  246  and a right channel signal to a right speaker  247 . Alternatively, the receiver  220  can be replaced by a television, a portable stereo system (e.g., a “boom box”), a clock-radio, and the like. The receiver  220  also provides the left and right channel signals to headphones  250 . A listener (user)  248  can listen to the left and right channel signals using the headphones  250  or the loudspeakers  246 ,  247 . The acoustic correction apparatus  120  can be implemented using analog devices in the receiver  220  or by software running on a Digital Signal Processor (DSP) in the receiver  220 . 
     The loudspeakers  246 ,  247  are often not optimally positioned to provide the user with the desired stereo image—thus decreasing the listening pleasure of a listener. In a similar manner, headphones, such as the headphones  250 , often produce a sound that is not pleasing because the headphones are placed adjacent to the ears rather than in front of the listener. Moreover, many small bookshelf loudspeakers, multimedia loudspeakers, and headphones have poor low frequency response characteristics that further decreasing the listening pleasure of the listener. The acoustic correction device (or software)  120  inside the receiver  220  corrects the left and right signals to produce a more pleasing sound when reproduced by the loudspeakers  246 ,  247  or the headphones  250 . In one embodiment, the receiver  220  includes controls (such as a width control  3846  shown in  FIG. 38  and/or a bass control  3827  shown in  FIG. 38 ) to allow the listener  248  to adjust the sound produced in the left and right channels according to whether the listener  248  is listening to the loudspeakers  246 ,  247  or the headphones  250 . 
       FIG. 3  illustrates a typical computer audio system  300  which may advantageously use an embodiment of the present invention to improve the audio performance produced by the loudspeakers  246 ,  247 . The loudspeakers  246 ,  247  are typically connected to a sound card (not shown) inside a computer unit  304 . The sound card can be any computer interface card that produces audio output, including a radio card, television tuner card, PCMCIA card, internal modem, plug-in Digital Signal Processor (DSP) card, etc. The computer  304  causes the sound card to generate audio signals that are converted by the loudspeakers  246  into acoustic waves. 
       FIG. 4A  depicts a graphical representation of a desired frequency response characteristic, appearing at the outer ears of a listener, within an audio reproduction environment. The curve  460  is a function of sound pressure level (SPL), measured in decibels, versus frequency. As can be seen in  FIG. 4A , the sound pressure level is relatively constant for all audible frequencies. The curve  460  can be achieved from reproduction of pink noise through a pair of ideal loudspeakers placed directly in front of a listener at approximately ear level. Pink noise refers to sound delivered over the audio frequency spectrum having equal energy per octave. In practice, the flat frequency response of the curve  460  may fluctuate in response to inherent acoustic limitations of speaker systems. 
     The curve  460  represents the sound pressure levels that exist before processing by the ear of a listener. Referring back to  FIG. 2 , the flat frequency response represented by the curve  460  is consistent with sound emanating towards the listener  248 , when the loudspeakers are located spaced apart and generally in front of the listener  248 . The human ear processes such sound, as represented by the curve  460 , by applying its own auditory response to the sound signals. This human auditory response is dictated by the outer pinna and the interior canal portions of the ear. 
     Unfortunately, the frequency response characteristics of many home and automotive sound reproduction systems do not provide the desired characteristic shown in  FIG. 4A . On the contrary, loudspeakers may be placed in acoustically-undesirable locations to accommodate other ergonomic requirements. Sound emanating from the loudspeakers  246  and  247  may be spectrally distorted by the mere placement of the loudspeakers  246  and  247  with respect to the listener  248 . Moreover, objects and surfaces in the listening environment may lead to absorption, or amplitude distortion, of the resulting sound signals. Such absorption is often prevalent among higher frequencies. 
     As a result of both spectral and amplitude distortion, a stereo image perceived by the listener  248  is spatially distorted providing an undesirable listening experience.  FIGS. 4B-4D  graphically depict levels of spatial distortion for various sound reproduction systems and listening environments. The distortion characteristics depicted in  FIGS. 4B-4D  represent sound pressure levels, measured in decibels, which are present near the ears of a listener. 
     The frequency response curve  464  of  FIG. 4B  has a decreasing sound-pressure level at frequencies above approximately 100 Hz. The curve  464  represents a possible sound pressure characteristic generated from loudspeakers, containing both woofers and tweeters, which are mounted below a listener. For example, assuming the loudspeakers  246  of  FIG. 2  contain tweeters, an audio signal played through only such loudspeakers  246  might exhibit the response of  FIG. 4B . 
     The particular slope associated with the decreasing curve  464  will vary, and may not be entirely linear, depending on the listening area, the quality of the loudspeakers, and the exact positioning of the loudspeakers within the listening area. For example, a listening environment with relatively hard surfaces will be more reflective of audio signals, particularly at higher frequencies, than a listening environment with relatively soft surfaces (e.g., cloth, carpet, acoustic tile, etc). The level of spectral distortion will vary as loudspeakers are placed further from, and positioned away from, a listener. 
       FIG. 4C  is a graphical representation of a sound-pressure versus frequency characteristic  468  wherein a first frequency range of audio signals are spectrally distorted, but a higher frequency range of the signals are not distorted. The characteristic curve  468  may be achieved from a speaker arrangement having low to mid-frequency loudspeakers placed below a listener and high-frequency loudspeakers positioned near, or at a listener&#39;s ear level. The sound image resulting from the characteristic curve  468  will have a low-frequency component positioned below the listener  248  of  FIG. 2 , and a high-frequency component positioned near the listener&#39;s ear level. 
       FIG. 4D  is a graphical representation of a sound-pressure versus frequency characteristic  470  having a reduced sound pressure level among lower frequencies and an increasing sound pressure level among higher frequencies. The characteristic  470  is achieved from a speaker arrangement having mid to low-frequency loudspeakers placed below a listener and high-frequency loudspeakers positioned above a listener. As the curve  470  of  FIG. 4D  indicates, the sound pressure level at frequencies above 1000 Hz may be significantly higher than lower frequencies, creating an undesirable audio effect for a nearby listener. The sound image resulting from the characteristic curve  470  will have a low-frequency component positioned below the listener  248  of  FIG. 2 , and a high-frequency component positioned above the listener  248 . 
     The audio characteristics of  FIGS. 4B-4D  represent various sound pressure levels obtainable in a common listening environment and heard by the listener  248 . The audio response curves of  FIGS. 4B-4D  are but a few examples of how audio signals present at the ears of a listener are distorted by various audio reproduction systems. The exact level of spatial distortion at any given frequency will vary widely depending on the reproduction system and the reproduction environment. The apparent location can be generated for a speaker system defined by apparent elevation and azimuth coordinates, with respect to a fixed listener, which are different from those of actual speaker locations. 
       FIG. 5  is block diagram of a stereo image correction system  122 , which inputs the left and right stereo signals  126  and  128 . The image-correction system  122  corrects the distorted spectral densities of various sound systems by advantageously dividing the audible frequency spectrum into a first frequency component, containing relatively lower frequencies, and a second frequency component, containing relatively higher frequencies. Each of the left and right signals  126  and  128  is separately processed through corresponding low-frequency correction systems  580 ,  582 , and high-frequency correction systems  584  and  586 . It should be pointed out that in one embodiment the correction systems  580  and  582  will operate in a relatively “low” frequency range of approximately 100 to 1000 Hertz, while the correction systems  584  and  586  will operate in a relatively “high” frequency range of approximately 1000 to 10,000 Hertz. This is not to be confused with the general audio terminology wherein low frequencies represent frequencies up to 100 Hertz, mid frequencies represent frequencies between 100 Hz to 4 kHz, and high frequencies represent frequencies above 4 kHz. 
     By separating the lower and higher frequency components of the input audio signals, corrections in sound pressure level can be made in one frequency range independent of the other. The correction systems  580 ,  582 ,  584 , and  586  modify the input signals  126  and  128  to correct for spectral and amplitude distortion of the input signals upon reproduction by loudspeakers. The resultant signals, along with the original input signals  126  and  128 , are combined at respective summing junctions  590  and  592 . The corrected left stereo signal, L c , and the corrected right stereo signal, R c , are provided along outputs to the bass enhancement unit  101 . 
     The corrected stereo signals provided to the bass unit  101  have a flat, i.e., uniform, frequency response appearing at the ears of the listener  248  (shown in  FIGS. 2 and 3 ). This spatially-corrected response creates an apparent source of sound which, when played through the loudspeakers  246  of  FIG. 2  or  3 , is seemingly positioned directly in front of the listener  248 . 
     Once the sound source is properly positioned through energy correction of the audio signal, the bass enhancement unit  101  corrects for low frequency deficiencies in the loudspeakers  246  and provides bass-corrected left and right channel signals to the stereo enhancement system  124 . The stereo enhancement system  124  conditions the stereo signals to broaden (horizontally) the stereo image emanating from the apparent sound source. As will be discussed in conjunction with  FIGS. 8A and 8B , the stereo image enhancement system  124  can be adjusted through a stereo orientation device to compensate for the actual location of the sound source. 
     In one embodiment, the stereo enhancement system  124  equalizes the difference signal information present in the left and right stereo signals 
     The left and right signals provided from the bass enhancement unit  101  are inputted by the enhancement system  124  and provided to a difference-signal generator  501  and a sum signal generator  504 . A difference signal (L c −R c ) representing the stereo content of the corrected left and right input signals, is presented at an output  502  of the difference signal generator  501 . A sum signal, (L c +R c ) representing the sum of the corrected left and right stereo signals is generated at an output  506  of the sum signal generator  504 . 
     The sum and difference signals at outputs  502  and  506  are provided to optional level-adjusting devices  508  and  510 , respectively. The devices  508  and  510  are typically potentiometers or similar variable-impedance devices. Adjustment of the devices  508  and  510  is typically performed manually to control the base level of sum and difference signal present in the output signals. This allows a user to tailor the level and aspect of stereo enhancement according to the type of sound reproduced, and depending on the user&#39;s personal preferences. An increase in the base level of the sum signal emphasizes the audio information at a center stage positioned between a pair of loudspeakers. Conversely, an increase in the base level of difference signal emphasizes the ambient sound information creating the perception of a wider sound image. In some audio arrangements where the music type and system configuration parameters are known, or where manual adjustment is not practical, the adjustment devices  508  and  510  may be eliminated requiring the sum and difference-signal levels to be predetermined and fixed. 
     The output of the device  510  is fed into a stereo enhancement equalizer  520  at an input  522 . The equalizer  520  spectrally shapes the difference signal appearing at the input  522  as shown in  FIG. 7  below. 
     The shaped difference signal is provided to a mixer  542 , which also receives the sum signal from the device  508 . In one embodiment, the stereo signals  594  and  596  are also provided to the mixer  542 . All of these signals are combined within the mixer  542  to produce an enhanced and spatially-corrected left output signal  530  and right output signal  532 . 
     Although the input signals  126  and  128  typically represent corrected stereo source signals, they may also be synthetically generated from a monophonic source. 
     Image Correction Characteristics 
       FIGS. 6A-6C  are graphical representations of the levels of spatial correction provided by “low” and “high”-frequency correction systems  580 ,  582 ,  584 ,  586  in order to obtain a relocated image generated from a pair of stereo signals. 
     Referring initially to  FIG. 6A , possible levels of spatial correction provided by the correction systems  580  and  582  are depicted as curves having different amplitude-versus-frequency characteristics. The maximum level of correction, or boost (measured in dB), provided by the systems  580  and  582  is represented by a correction curve  650 . The curve  650  provides an increasing level of boost within a first frequency range of approximately 100 Hz and 1000 Hz. At frequencies above 1000 Hz, the level of boost is maintained at a fairly constant level. A curve  652  represents a near-zero level of correction. 
     To those skilled in the art, a typical filter is usually characterized by a pass-band and stop-band of frequencies separated by a cutoff frequency. The correction curves, of  FIGS. 6A-6C , although representative of typical signal filters, can be characterized by a pass-band, a stop-band, and a transition band. A filter constructed in accordance with the characteristics of  FIG. 6A  has a pass-band above approximately 1000 Hz, a transition-band between approximately 100 and 1000 Hz, and a stop-band below approximately 100 Hz. Filters according to  FIG. 6B  have pass-bands above approximately 10 kHz, transition-bands between approximately 1 kHz and 10 kHz, and a stop-band below approximately 1 kHz. Filters according to  FIG. 6C  have stop-bands above approximately 10 kHz, transition-bands between approximately 1 kHz and 10 kHz, and pass-bands below approximately 1 kHz. In one embodiment, the filters are first-order filters. 
     As can be seen in  FIGS. 6A-6C , spatial correction of an audio signal by the systems  580 ,  582 ,  584 , and  586  is substantially uniform within the pass-bands, but is largely frequency-dependent within the transition bands. The amount of acoustic correction applied to an audio signal can be varied as a function of frequency through adjustment of the stereo image correction system  122 , which varies the slope of the transition bands of  FIGS. 6A-6C . As a result, frequency-dependent correction is applied to a first frequency range between 100 and 1000 hertz, and applied to a second frequency range of 1000 to 10,000 hertz. An infinite number of correction curves are possible through independent adjustment of the correction systems  580 ,  582 ,  584  and  586 . 
     In accordance with one embodiment, spatial correction of the higher frequency stereo-signal components occurs between approximately 1000 Hz and 10,000 Hz. Energy correction of these signal components may be positive, i.e., boosted, as depicted in  FIG. 6B , or negative, i.e., attenuated, as depicted in  FIG. 6C . The range of boost provided by the correction systems  584 ,  586  is characterized by a maximum-boost curve  660  and a minimum-boost curve  662 . Curves  664 ,  666 , and  668  represent still other levels of boost, which may be required to spatially correct sound emanating from different sound reproduction systems.  FIG. 6C  depicts energy-correction curves that are essentially the inverse of those in  FIG. 6B . 
     Since the lower frequency and higher frequency correction factors, represented by the curves of  FIGS. 6A-6C , are added together, there is a wide range of possible spatial correction curves applicable between the frequencies of 100 to 10,000 Hz.  FIG. 6D  is a graphical representation depicting a range of composite spatial correction characteristics provided by the stereo image correction system  122 . Specifically, the solid line curve  680  represents a maximum level of spatial correction comprised of the curve  650  (shown in  FIG. 6A ) and the curve  660  (shown in  FIG. 6B ). Correction of the lower frequencies may vary from the solid curve  680  through the range designated by θ 1 . Similarly, correction of the higher frequencies may vary from the solid curve  680  through the range designated by θ 2 . Accordingly, the amount of boost applied to the first frequency range of 100 to 1000 Hertz varies between approximately 0 and 15 dB, while the correction applied to the second frequency range of 1000 to 10,000 Hertz may vary from approximately 13 dB to −15 dB. 
     Image Enhancement Characteristics 
     Turning now to the stereo image enhancement aspect of the present invention, a series of perspective-enhancement, or normalization curves, is graphically represented in  FIG. 7 . The signal (L c −R c ) p  represents the processed difference signal which has been spectrally shaped according to the frequency-response characteristics of  FIG. 7 . These frequency-response characteristics are applied by the equalizer  520  depicted in  FIG. 5  and are partially based upon HRTF principles. 
     In general, selective amplification of the difference signal enhances any ambient or reverberant sound effects which may be present in the difference signal but which are masked by more intense direct-field sounds. These ambient sounds are readily perceived in a live sound stage at the appropriate level. In a recorded performance, however, the ambient sounds are attenuated relative to a live performance. By boosting the level of difference signal derived from a pair of stereo left and right signals, a projected sound image can be broadened significantly when the image emanates from a pair of loudspeakers placed in front of a listener. 
     The perspective curves  790 ,  792 ,  794 ,  796 , and  798  of  FIG. 7  are displayed as a function of gain against audible frequencies displayed in log format. The different levels of equalization between the curves of  FIG. 7  are required to account for various audio reproduction systems. In one embodiment, the level of difference-signal equalization is a function of the actual placement of loudspeakers relative to a listener within an audio reproduction system. The curves  790 ,  792 ,  794 ,  796 , and  798  generally display a frequency contouring characteristic wherein lower and higher difference-signal frequencies are boosted relative to a mid-band of frequencies. 
     According to one embodiment, the range for the perspective curves of  FIG. 7  is defined by a maximum gain of approximately 10-15 dB located at approximately 125 to 150 Hz. The maximum gain values denote a turning point for the curves of  FIG. 7  whereby the slopes of the curves  790 ,  792 ,  794 ,  796 , and  798  change from a positive value to a negative value. Such turning points are labeled as points A, B, C, D, and E in  FIG. 7 . The gain of the perspective curves decreases below 125 Hz at a rate of approximately 6 dB per octave. Above 125 Hz, the gain of the curves of  FIG. 7  also decreases, but at variable rates, towards a minimum-gain turning point of approximately −2 to +10 dB. The minimum-gain turning points vary significantly between the curves  790 ,  792 ,  794 ,  796 , and  798 . The minimum-gain turning points are labeled as points A′, B′, C′, D′, and E′, respectively. The frequencies at which the minimum-gain turning points occur varies from approximately 2.1 kHz for curve  790  to approximately 5 kHz for curve  798 . The gain of the curves  790 ,  792 ,  794 ,  796 , and  798  increases above their respective minimum-gain frequencies up to approximately 10 kHz. Above 10 kHz, the gain applied by the perspective curves begins to level off. An increase in gain will continue to be applied by all of the curves, however, up to approximately 20 kHz, i.e., approximately the highest frequency audible to the human ear. 
     The preceding gain and frequency figures are merely design objectives and the actual figures will likely vary from system to system. Moreover, adjustment of the signal level devices  508  and  510  will affect the maximum and minimum gain values, as well as the gain separation between the maximum-gain frequency and the minimum-gain frequency. 
     Equalization of the difference signal in accordance with the curves of  FIG. 7  is intended to boost the difference signal components of statistically lower intensity without overemphasizing the higher-intensity difference signal components. The higher-intensity difference signal components of a typical stereo signal are found in a mid-range of frequencies between approximately 1 to 4 kHz. The human ear has a heightened sensitivity to these same mid-range of frequencies. Accordingly, the enhanced left and right output signals  530  and  532  produce a much improved audio effect because ambient sounds are selectively emphasized to fully encompass a listener within a reproduced sound stage. 
     As can be seen in  FIG. 7 , difference signal frequencies below 125 Hz receive a decreased amount of boost, if any, through the application of the perspective curve. This decrease is intended to avoid over-amplification of very low, i.e., bass, frequencies. With many audio reproduction systems, amplifying an audio difference signal in this low-frequency range can create an unpleasurable and unrealistic sound image having too much bass response. Examples of such audio reproduction systems include near-field or low-power audio systems, such as multimedia computer systems, as well as home stereo systems. A large draw of power in these systems may cause amplifier “clipping” during periods of high boost, or it may damage components of the audio system including the loudspeakers. Limiting the bass response of the difference signal also helps avoid these problems in most near-field audio enhancement applications. 
     In accordance with one embodiment, the level of difference signal equalization in an audio environment having a stationary listener is dependent upon the actual speaker types and their locations with respect to the listener. The acoustic principles underlying this determination can best be described in conjunction with  FIGS. 8A and 8B .  FIGS. 8A and 8B  are intended to show such acoustic principles with respect to changes in azimuth of a speaker system. 
       FIG. 8A  depicts a top view of a sound reproduction environment having loudspeakers  800  and  802  placed slightly forward of, and pointed towards, the sides of a listener  804 . The loudspeakers  800  and  802  are also placed below the listener  804  at an elevational position similar to that of the loudspeakers  246  shown in  FIG. 2 . Reference planes A and B are aligned with ears  806 ,  808  of the listener  804 . The planes A and B are parallel to the listener&#39;s line-of-sight as shown. 
     The location of the loudspeakers preferably correspond to the locations of the loudspeakers  810  and  812 . In one embodiment, when the loudspeakers cannot be located in a desired position, enhancement of the apparent sound image can be accomplished by selectively equalizing the difference signal, i.e., the gain of the difference signal will vary with frequency. The curve  790  of  FIG. 7  represents the desired level of difference-signal equalization with actual speaker locations corresponding to the phantom loudspeakers  810  and  812 . 
     Bass Enhancement 
     The present invention also provides a method and system for enhancing audio signals. The sound enhancement system improves the realism of sound with a unique sound enhancement process. Generally speaking, the sound enhancement process receives two input signals, a left input signal and a right input signal, and in turn, generates two enhanced output signals, a left output signal, and a right output signal. 
     The left and right input signals are processed collectively to provide a pair of left and right output signals. In particular, the enhanced system embodiment equalizes the differences that exist between the two input signals in a manner which broadens and enhances the perceived bandwidth of the sounds. In addition, many embodiments adjust the level of the sound that is common to both input signals so as to reduce clipping. Advantageously, some embodiments achieve sound enhancement with simplified, low cost, and easy-to-manufacture analog systems that do not require digital signal processing. 
     Although the embodiments are described herein with reference to one sound enhancement systems, the invention is not so limited, and can be used in a variety of other contexts in which it is desirable to adapt different embodiments of the sound enhancement system to different situations. 
     A typical small loudspeaker system used for multimedia computers, automobiles, small stereophonic systems, portable stereophonic systems, headphones, and the like, will have an acoustic output response that rolls off at about 150 Hz.  FIG. 9  shows a curve  906  corresponding approximately to the frequency response of the human ear.  FIG. 9  also shows the measured response  908  of a typical small computer loudspeaker system that uses a high-frequency driver (tweeter) to reproduce the high frequencies, and a four inch midrange-bass driver (woofer) to reproduce the midrange and bass frequencies. Such a system employing two drivers is often called a two-way system. Loudspeaker systems employing more than two drivers are known in the art and will work with an embodiment of the present invention. Loudspeaker systems with a single driver are also known and will also work with the present invention. The response  908  is plotted on a rectangular plot with an X-axis showing frequencies from 20 Hz to 20 kHz. This frequency band corresponds to the range of normal human hearing. The Y-axis in  FIG. 9  shows normalized amplitude response from 0 dB to −50 dB. The curve  908  is relatively flat in a midrange frequency band from approximately 2 kHz to 10 kHz, showing some rolloff above 10 kHz. In the low frequency ranges, the curve  908  exhibits a low-frequency rolloff that begins in a midbass band between approximately 150 Hz and 2 kHz such that below 150 Hz, the loudspeaker system produces very little acoustic output. 
     The location of the frequency bands shown in  FIG. 9  are used by way of example and not by way of limitation. The actual frequency ranges of the deep bass band, midbass band, and midrange band vary according to the loudspeaker and the application for which the loudspeaker is used. The term deep bass is used, generally, to refer to frequencies in a band where the loudspeaker produces an output that is less accurate as compared to the loudspeaker output at higher frequencies, such as, for example, in the midbass band. The term midbass band is used, generally, to refer to frequencies above the deep bass band. The term midrange is used, generally, to refer to frequencies above the midbass band. 
     Many cone-type drivers are very inefficient when producing acoustic energy at low frequencies where the diameter of the cone is less than the wavelength of the acoustic sound wave. When the cone diameter is smaller than the wavelength, maintaining a uniform sound pressure level of acoustic output from the cone requires that the cone excursion be increased by a factor of four for each octave (factor of 2) that the frequency drops. The maximum allowable cone excursion of the driver is quickly reached if one attempts to improve low-frequency response by simply boosting the electrical power supplied to the driver. 
     Thus, the low-frequency output of a driver cannot be increased beyond a certain limit, and this explains the poor low-frequency sound quality of most small loudspeaker systems. The curve  908  is typical of most small loudspeaker systems that employ a low-frequency driver of approximately four inches in diameter. Loudspeaker systems with larger drivers will tend to produce appreciable acoustic output down to frequencies somewhat lower than those shown in the curve  908 , and systems with smaller low-frequency drivers will typically not produce output as low as that shown in the curve  908 . 
     As discussed above, to date, a system designer has had little choice when designing loudspeaker systems with extended low-frequency response. Previously known solutions were expensive and produced loudspeakers that were too large for the desktop. One popular solution to the low-frequency problem is the use of a sub-woofer, which is usually placed on the floor near the computer system. Sub-woofers can provide adequate low-frequency output, but they are expensive, and thus relatively uncommon as compared to inexpensive desktop loudspeakers. 
     Rather than use drivers with large diameter cones, or a sub-woofer, an embodiment of the present invention overcomes the low-frequency limitations of small systems by using characteristics of the human hearing system to produce the perception of low-frequency acoustic energy, even when such energy is not produced by the loudspeaker system. 
     The human auditory system is known to be non-linear. A non-linear system is, simply put, a system where an increase in the input is not followed by a proportional increase in the output. Thus, for example, in the ear, a doubling of the acoustic sound pressure level does not produce a perception that the volume of the sound source has been doubled. In fact, the human ear is, to a first approximation, a square-law device that is responsive to power rather than intensity of the acoustic energy. This non-linearity of the hearing mechanism produces intermodulation frequencies that are heard as overtones or harmonics of the actual frequencies in the acoustic wave. 
     The intermodulation effect of the non-linearities in the human ear is shown in  FIG. 10 , which illustrates an idealized amplitude spectrum of two pure tones. The spectral diagram in  FIG. 10  shows a first spectral line  1004  which corresponds to acoustic energy produced by a loudspeaker driver (e.g., a sub-woofer) at 50 Hz. A second spectral line  1002  is shown at 60 Hz. The lines  1004  and  1002  are actual spectral lines corresponding to real acoustic energy produced by the driver, and no other acoustic energy is assumed to exist. Nevertheless, the human ear, because of its inherent non-linearities, will produce intermodulation products corresponding to the sum of the two actual spectral frequencies and the difference between the two spectral frequencies. 
     For example, a person listening to the acoustic energy represented by the spectral lines  1004  and  1002  will perceive acoustic energy at 50 Hz, as shown by the spectral line  1006 , at 60 Hz, as shown by the spectral line  1008 , and at 110 Hz, as shown by the spectra line  1010 . The spectral line  1010  does not correspond to real acoustic energy produced by the loudspeaker, but rather corresponds to a spectral line created inside the ear by the non-linearities of the ear. The line  1010  occurs at a frequency of 110 Hz which is the sum of the two actual spectral lines (110 Hz=50 Hz+60 Hz). Note that the non-linearities of the ear will also create a spectral line at the difference frequency of 10 Hz (10 Hz=60 Hz−50 Hz), but that line is not perceived because it is below the range of human hearing. 
       FIG. 10  illustrates the process of intermodulation inside the human ear, but it is somewhat simplified when compared to real program material, such as music. Typical program material such as music is rich in harmonics, so much so that most music exhibits an almost continuous spectrum, as shown in  FIG. 11 .  FIG. 11  shows the same type of comparison between actual and perceived acoustic energy, as shown in  FIG. 10 , except that the curves in  FIG. 11  are shown for continuous spectra.  FIG. 11  shows an actual acoustic energy curve  1120  and the corresponding perceived spectrum  1130 . 
     As with most non-linear systems, the non-linearity of the ear is more pronounced when the system is making large excursions (e.g., large signal levels) than for small excursions. Thus, for the human ear, the non-linearities are more pronounced at low frequencies, where the eardrum and other elements of the ear make relatively large mechanical excursions, even at lower volume levels. Thus,  FIG. 11  shows that the difference between actual acoustic energy  1120 , and the perceived acoustic energy  1130  tends to be greatest in the lower-frequency range and becomes relatively smaller at the higher-frequency range. 
     As shown in  FIGS. 10 and 11 , low-frequency acoustic energy comprising multiple tones or frequencies will produce, in the listener, the perception that the acoustic energy in the midbass range contains more spectral content than actually exists. The human brain, when faced with a situation where information is thought to be missing, will attempt to “fill in” missing information on a subconscious level. This filling in phenomenon is the basis for many optical illusions. In an embodiment of the present invention, the brain can be tricked into filling in low-frequency information that is not really present by providing the brain with the midbass effects of such low-frequency information. 
     In other words, if the brain is presented with the harmonics that would be produced by the ear if the low-frequency acoustic energy was present (e.g., the spectral line  1010 ) then under the right conditions, the brain will subconsciously fill in the low-frequency spectral lines  1006  and  1008  which it thinks “must” be present. This filling in process is augmented by another effect of the non-linearity of the human ear known as the detector effect. 
     The non-linearity of the human ear also causes the ear to act like a detector, similar to a diode detector in an Amplitude Modulation (AM) receiver. If a midbass harmonic tone is AM modulated by a deep bass tone, the ear will demodulate the modulated midbass carrier to reproduce the deep bass envelope.  FIGS. 12A and 12B  graphically illustrate the modulated and demodulated signal.  FIG. 12A  shows, on a time axis, a modulated signal comprising a higher-frequency carrier signal (e.g. the midbass carrier) modulated by a deep bass signal. 
     The amplitude of the higher-frequency signal is modulated by a lower frequency tone, and thus, the amplitude of the higher-frequency signal varies according to the frequency of the lower frequency tone. The non-linearity of the ear will partially demodulate the signal such that the ear will detect the low-frequency envelope of the higher-frequency signal, and thus produce the perception of the low-frequency tone, even though no actual acoustic energy was produced at the lower frequency. As with the intermodulation effect discussed above, the detector effect can be enhanced by proper signal processing of the signals in the midbass frequency range. By using the proper signal processing, it is possible to design a sound enhancement system that produces the perception of low-frequency acoustic energy, even when using loudspeakers that are incapable of, or inefficient at, producing such energy. 
     The perception of the actual frequencies present in the acoustic energy produced by the loudspeaker may be deemed a first order effect. The perception of additional harmonics not present in the actual acoustic frequencies, whether such harmonics are produced by intermodulation distortion or detection, may be deemed a second order effect. 
     Bass Enhancement Expander 
       FIG. 13A  is a block diagram of a sound system wherein the sound enhancement function is provided by a bass enhancement unit  1304 . The bass enhancement unit  1304  receives audio signals from a signal source  1302 . The signal source  1302  may be any signal source, including the signal processing block  122  shown in  FIG. 1 . The bass enhancement unit  1304  performs signal processing to modify the received audio signals to produce audio output signals. The audio output signals may be provided to loudspeakers, amplifiers, or other signal processing devices. 
       FIG. 13B  is a block diagram of a topology for a two-channel bass enhancement unit  1304  having a first input  1309 , a second input  1311 , a first output  1317 , and a second output  1319 . The first input  1309  and first output  1317  correspond to a first channel. The second input  1311  and second output  1319  correspond to a second channel. The first input  1309  is provided to a first input of a combiner  1310  and to an input of a signal processing block  1313 . An output of the signal processing block  1313  is provided to a first input of a combiner  1314 . The second input  1311  is provided to a second input of the combiner  1310  and to an input of a signal processing block  1315 . An output of the signal processing block  1315  is provided to a first input of a combiner  1316 . An output of the combiner  1310  is provided to an input of a signal processing block  1312 . An output of the signal processing block  1312  is provided to a second input of the combiner  1314  and to a second input of the combiner  1316 . An output of the combiner  1314  is provided to the first output  1317 . An output of the second combiner  1316  is provided to the second output  1319 . 
     Signals from the first and second inputs  1309  and  1311  are combined and processed by the signal processing block  1312 . The output of the signal processing block  1312  is a signal, that when combined with the outputs of the signal processing blocks  1313  and  1315 , respectively, produces the bass enhanced outputs  1317  and  1319 . 
       FIG. 13C  is a block diagram of another topology for a two-channel bass enhancement unit  1344 . In  FIG. 13C , the first input  1309  is provided to an input of a signal processing block  1321  and to an input of a signal processing block  1322 . An output of the signal processing block  1321  is provided to a first input of a combiner  1325  and an output of the signal processing block  1322  is provided to a second input of the combiner  1325 . The second input  1311  is provided to an input of a signal processing block  1323  and to an input of a signal processing block  1324 . An output of the signal processing block  1323  is provided to a first input of a combiner  1326  and an output of the signal processing block  1324  is provided to a second input of the combiner  1326 . An output of the combiner  1325  is provided to the first output  1317  and an output of the second combiner  1326  is provided to the second output  1319 . 
     Unlike the topology shown in  FIG. 13B , the topology shown in  FIG. 13C  does not combine the two input signals  1309  and  1311 , but, rather, the two channels are kept separate, and the bass enhancement processing is performed on each channel. 
       FIG. 14  is a block diagram  1400  of one embodiment of the bass enhancement system  1304  shown in  FIG. 13A . The bass enhancement system  1400  uses a bass punch unit  1420  to generate a time-dependent enhancement factor.  FIG. 14  may also be used as a flowchart to describe a program running on a DSP or other processor which implements the signal processing operations of an embodiment of the present invention.  FIG. 14  shows two inputs, a left-channel input  1402  and a right-channel input  1404 . As with previous embodiments, left and right are used as a convenience, not as a limitation. The inputs  1402  and  1404  are both provided to an adder  1406  that produces an output that is a combination of the two inputs. 
     The output of the adder  1406  is provided to a first bandpass filter  1412 , a second bandpass filter  1413 , a third bandpass filter  1415 , and a fourth bandpass filter  1411 . The output of the bandpass filter  1413  is provided to an input of an adder  1418 . 
     The output of the bandpass filter  1415  is provided to a first throw of a single pole double throw (SPDT) switch  1416 . The output of the bandpass filter  1411  is provided to a second throw of the SPDT switch  1416 . The pole of the switch  1416  is provided to an input of the adder  1418 . 
     The output of the bandpass filter  1412  is provided to an input of the adder  1418 . 
     An output of the adder  1418  is provided to an input of the bass punch unit  1420 . An output of the bass punch unit  1420  is provided to a first throw of a (SPDT) switch  1422 . A second throw of the SPDT switch  1422  is provided to ground. A pole of the SPDT switch  1422  is provided to a first input of a left-channel adder  1424  and to a first input of a right-channel adder  1432 . The left-channel input  1402  is provided to a second input of the left-channel adder  1424  and the right-channel input  1404  is provided to a second input of the right-channel adder  1432 . The outputs of the left-channel adder  1424  and the right-channel adder  1432  are, respectively, a left-channel output  1430  and a right-channel output  1433  of the signal processing block  1400 . The switches  1422  and  1416  are optional and may be replaced by fixed connections. 
     The switch  1416  allows the filters  1411 - 1415  to be configured for two different frequency ranges, namely 40-150 Hz, and 100-200 Hz. 
     The filtering operations provided by the filters  1411 - 1413 ,  1415  and the combiner  1418  may be combined into a composite filter  1407  as shown in  FIG. 14 . For example, in an alternative embodiment, the filters  1411 - 1413 ,  1415  are combined into a single bandpass filter having a passband that extends from approximately 40 Hz to 250 Hz. For processing bass frequencies, the passband of the composite filter  1407  preferably extends from approximately 20 to 100 Hz at the low-end, and from approximately 150 to 350 Hz at the high-end. The composite filter  1407  may have other filter transfer functions as well, including, for example, a highpass filter, a shelving filter, etc. The composite filter may also be configured to operate in a manner similar to a graphic equalizer and attenuate some frequencies within its passband relative to other frequencies within its passband. 
     As shown,  FIG. 14  corresponds approximately to the topology shown in  FIG. 13B , where the signal processing blocks  1313  and  1315  have a transfer function of unity and the signal processing block  1312  comprises the composite filter  1407  and the bass punch unit  1420 . However, the signal processing shown in  FIG. 14  is not limited to the topology shown in  FIG. 13B . The elements of  FIG. 14  may also be used in the topology shown in  FIG. 13C , where the signal processing blocks  1321  and  1323  have a transfer function of unity and the signal processing blocks  1322  and  1324  comprise the composite filter  1407  and the bass punch unit  1420 . Although not shown in  FIG. 14 , the signal processing blocks  1313 ,  1315 ,  1321 , and  1323  may provide additional signal processing, such as, for example, high pass filtering to remove low bass frequencies, high pass filtering to remove frequencies processed by the bass punch unit  1420 , high frequency emphasis to enhance the high frequency sounds, additional mid bass processing to supplement the bass punch system, etc. Other combinations are contemplated as well. 
       FIG. 15  is a frequency-domain plot that shows the general shape of the transfer functions of the bandpass filters  1411 - 1413 ,  1415 .  FIG. 15  shows the bandpass transfer functions  1501 - 1504 , corresponding to the bandpass filters  1411 - 1413 ,  1415  respectively. The transfer functions  1501 - 1504  are shown as bandpass functions centered at 40, 100, 150, and 200 Hz respectively. 
     In one embodiment, the bandpass filter  1411  is tuned to a frequency below 100 Hz, such as 40 Hz. When the switch  1416  is in a first position, corresponding to the first throw, it selects the bandpass filter  1411  and deselects the bandpass filter  1415 , thereby providing bandpass filters at 40, 100, and 150 Hz. When the switch  1416  is in a second position, corresponding to the second throw, it deselects the bandpass filter  1411  and selects the bandpass filter  1415 , thus providing bandpass filters at 100, 150, and 200 Hz. 
     Thus, the switch  1416  desirably allows a user to select the frequency range to be enhanced. A user with a loudspeaker system that provides small woofers, such as woofer of three to four inches in diameter, will typically select the upper frequency range provided by the bandpass filters  1412 - 1413 ,  1415 , which are tuned to 100, 150, and 200 Hz respectively. A user with a loudspeaker system that provides somewhat larger woofers, such as woofers of approximately five inches in diameter or larger, will typically select the lower frequency range provided by the bandpass filters  1411 - 1413 , which are tuned to 40, 100, and 150 Hz respectively. One skilled in the art will recognize that more switches could be provided to allow selection of more bandpass filters and more frequency ranges. Selecting different bandpass filters to provide different frequency ranges is a desirable technique because the bandpass filters are inexpensive and because different bandpass filters can be selected with a single-throw switch. 
     In one embodiment, the bass punch unit  1420  uses an Automatic Gain Control (AGC) comprising a linear amplifier with an internal servo feedback loop. The servo automatically adjusts the average amplitude of the output signal to match the average amplitude of a signal on the control input. The average amplitude of the control input is typically obtained by detecting the envelope of the control signal. The control signal may also be obtained by other methods, including, for example, lowpass filtering, bandpass filtering, peak detection, RMS averaging, mean value averaging, etc. 
     In response to an increase in the amplitude of the envelope of the signal provided to the input of the bass punch unit  1420 , the servo loop increases the forward gain of the bass punch unit  1420 . Conversely, in response to a decrease in the amplitude of the envelope of the signal provided to the input of the bass punch unit  1420 , the servo loop decreases the forward gain of the bass punch unit  1420 . In one embodiment, the gain of the bass punch unit  1420  increases more rapidly that the gain decreases.  FIG. 16  is a time domain plot that illustrates the gain of the bass punch unit  1420  in response to a unit step input. One skilled in the art will recognize that  FIG. 16  is a plot of gain as a function of time, rather than an output signal as a function of time. Most amplifiers have a gain that is fixed, so gain is rarely plotted. However, the Automatic Gain Control (AGC) in the bass punch unit  1420  varies the gain of the bass punch unit  1420  in response to the envelope of the input signal. 
     The unit step input is plotted as a curve  1609  and the gain is plotted as a curve  1602 . In response to the leading edge of the input pulse  1609 , the gain rises during a period  1604  corresponding to an attack time constant. At the end of the time period  1604 , the gain  1602  reaches a steady-state gain of A 0 . In response to the trailing edge of the input pulse  1609 , the gain falls back to zero during a period corresponding to a decay time constant  1606 . 
     The attack time constant  1604  and the decay time constant  1606  are desirably selected to provide enhancement of the bass frequencies without overdriving other components of the system such as the amplifier and loudspeakers.  FIG. 17  is a time-domain plot  1700  of a typical bass note played by a musical instrument such as a bass guitar, bass drum, synthesizer, etc. The plot  1700  shows a higher-frequency portion  1744  that is amplitude modulated by a lower-frequency portion having a modulation envelope  1742 . The envelope  1742  has an attack portion  1746 , followed by a decay portion  1747 , followed by a sustain portion  1748 , and finally, followed by a release portion  1749 . The largest amplitude of the plot  1700  is at a peak  1750 , which occurs at the point in time between the attack portion  1746  and the decay portion  1747 . 
     As stated, the waveform  1744  is typical of many, if not most, musical instruments. For example, a guitar string, when pulled and released, will initially make a few large amplitude vibrations, and then settle down into a more or less steady state vibration that slowly decays over a long period. The initial large excursion vibrations of the guitar string correspond to the attack portion  1746  and the decay portion  1747 . The slowly decaying vibrations correspond to the sustain portion  1748  and the release portions  1749 . Piano strings operate in a similar fashion when struck by a hammer attached to a piano key. 
     Piano strings may have a more pronounced transition from the sustain portion  1748  to the release portion  1749 , because the hammer does not return to rest on the string until the piano key is released. While the piano key is held down, during the sustain period  1748 , the string vibrates freely with relatively little attenuation. When the key is released, the felt covered hammer comes to rest on the key and rapidly damps out the vibration of the string during the release period  1749 . 
     Similarly, a drumhead, when struck, will produce an initial set of large excursion vibrations corresponding to the attack portion  1746  and the decay portion  1747 . After the large excursion vibrations have died down (corresponding to the end of the decay portion  1747 ) the drumhead will continue to vibrate for a period of time corresponding to the sustain portion  1748  and release portion  1749 . Many musical instrument sounds can be created merely by controlling the length of the periods  1746 - 1749 . 
     As described in connection with  FIG. 12A , the amplitude of the higher-frequency signal is modulated by a lower-frequency tone (the envelope), and thus, the amplitude of the higher-frequency signal varies according to the frequency of the lower frequency tone. The non-linearity of the ear will partially demodulate the signal such that the ear will detect the low-frequency envelope of the higher-frequency signal, and thus produce the perception of the low-frequency tone, even though no actual acoustic energy was produced at the lower frequency. The detector effect can be enhanced by proper signal processing of the signals in the midbass frequency range, typically between 50-150 Hz on the low end of the range and 200-500 Hz on the high end of the range. By using the proper signal processing, it is possible to design a sound enhancement system that produces the perception of low-frequency acoustic energy, even when using loudspeakers that are incapable of producing such energy. 
     The perception of the actual frequencies present in the acoustic energy produced by the loudspeaker may be deemed a first order effect. The perception of additional harmonics not present in the actual acoustic frequencies, whether such harmonics are produced by intermodulation distortion or detection may be deemed a second order effect. 
     However, if the amplitude of the peak  1750  is too high, the loudspeakers (and possibly the power amplifier) will be overdriven. Overdriving the loudspeakers will cause a considerable distortion and may damage the loudspeakers. 
     The bass punch unit  1420  desirably provides enhanced bass in the midbass region while reducing the overdrive effects of the peak  1750 . The attack time constant  1604  provided by the bass punch unit  1420  limits the rise time of the gain through the bass punch unit  1420 . The attack time constant of the bass punch unit  1420  has relatively less effect on a waveform with a long attack period  1746  (slow envelope risetime) and relatively more effect on a waveform with a short attack period  1746  (fast envelope risetime). 
     Bass Punch with Peak Compression 
     An attack portion of a note played by a bass instrument (e.g., a bass guitar) will often begin with an initial pulse of relatively high amplitude. This peak may, in some cases, overdrive the amplifier or loudspeaker causing distorted sound and possibly damaging the loudspeaker or amplifier. The bass enhancement processor provides a flattening of the peaks in the bass signal while increasing the energy in the bass signal, thereby increasing the overall perception of bass. 
     The energy in a signal is a function of the amplitude of the signal and the duration of the signal. Stated differently, the energy is proportional to the area under the envelope of the signal. Although the initial pulse of a bass note may have a relatively large amplitude, the pulse often contains little energy because it is of short duration. Thus, the initial pulse, having little energy, often does not contribute significantly to the perception of bass. Accordingly, the initial pulse can usually be reduced in amplitude without significantly affecting the perception of bass. 
       FIG. 18  is a signal processing block diagram of a bass enhancement system  1800  that provides bass enhancement using a peak compressor to control the amplitude of pulses, such as the initial pulse, bass notes. In the system  1800 , a peak compressor  1802  is interposed between the combiner  1718  and the punch unit  1720 . The output of the combiner  1718  is provided to an input of the peak compressor  1802 , and an output of the peak compressor  1802  is provided to the input of the bass punch unit  1720 . 
     The comments above relating  FIG. 14  to  FIGS. 13B and 13C  apply to the topology shown in  FIG. 18  as well. For example, as shown,  FIG. 18  corresponds approximately to the topology shown in  FIG. 13B , where the signal processing blocks  1313  and  1315  have a transfer function of unity and the signal processing block  1312  comprises the composite filter  1707 , the peak compressor  1802 , and the bass punch unit  1720 . However, the signal processing shown in  FIG. 18  is not limited to the topology shown in  FIG. 13B . The elements of  FIG. 18  may also be used in the topology shown in  FIG. 13C . Although not shown in  FIG. 18 , the signal processing blocks  1313 ,  1315 ,  1321 , and  1323  may provide additional signal processing, such as, for example, high pass filtering to remove low bass frequencies, high pass filtering to remove frequencies processed by the bass punch unit  1702  and the compressor  1802 , high frequency emphasis to enhance the high frequency sounds, additional mid bass processing to supplement the bass punch system  1720  and peak compressor  1802 , etc. Other combinations are contemplated as well. 
     The peak compression unit  1802  “flattens” the envelope of the signal provided at its input. For input signals with a large amplitude, the apparent gain of the compression unit  1802  is reduced. For input signals with a small amplitude, the apparent gain of the compression unit  1802  is increased. Thus, the compression unit reduces the peaks of the envelope of the input signal (and fills in the troughs in the envelope of the input signal). Regardless of the signal provided at the input of the compression unit  1802 , the envelope (e.g., the average amplitude) of the output signal from the compression unit  1802  has a relatively uniform amplitude. 
       FIG. 19  is a time-domain plot showing the effect of the peak compressor on an envelope with an initial pulse of relatively high amplitude.  FIG. 19  shows a time-domain plot of an input envelope  1914  having an initial large amplitude pulse followed by a longer period of lower amplitude signal. An output envelope  1916  shows the effect of the bass punch unit  1720  on the input envelope  1914  (without the peak compressor  1802 ). An output envelope  1917  shows the effect of passing the input signal  1914  through both the peak compressor  1802  and the punch unit  1720 . 
     As shown in  FIG. 19 , assuming the amplitude of the input signal  1914  is sufficient to overdrive the amplifier or loudspeaker, the bass punch unit does not limit the maximum amplitude of the input signal  1914  and thus the output signal  1916  is also sufficient to overdrive the amplifier or loudspeaker. 
     The pulse compression unit  1802  used in connection with the signal  1917 , however, compresses (reduces the amplitude of) large amplitude pulses. The compression unit  1802  detects the large amplitude excursion of the input signal  1914  and compresses (reduces) the maximum amplitude so that the output signal  1917  is less likely to overdrive the amplifier or loudspeaker. 
     Since the compression unit  1802  reduces the maximum amplitude of the signal, it is possible to increase the gain provided by the punch unit  1420  without significantly reducing the probability that the output signal  1917  will overdrive the amplifier or loudspeaker. The signal  1917  corresponds to an embodiment where the gain of the bass punch unit  1420  has been increased. Thus, during the long decay portion, the signal  1917  has a larger amplitude than the curve  1916 . 
     As described above, the energy in the signals  1914 ,  1916 , and  1917  is proportional to the area under the curve representing each signal. The signal  1917  has more energy because, even though it has a smaller maximum amplitude, there is more area under the curve representing the signal  1917  than either of the signals  1914  or  1916 . Since the signal  1917  contains more energy, a listener will perceive more bass in the signal  1917 . 
     Thus, the use of the peak compressor in combination with the bass punch unit  1420  allows the bass enhancement system to provide more energy in the bass signal, while reducing the likelihood that the enhanced bass signal will overdrive the amplifier or loudspeaker. 
     Stereo Image Enhancement 
     The present invention also provides a method and system that improves the realism of sound (especially the horizontal aspects of the sound stage) with a unique differential perspective correction system. Generally speaking, the differential perspective correction apparatus receives two input signals, a left input signal and a right input signal, and in turn, generates two enhanced output signals, a left output signal and a right output signal as shown in connection with  FIG. 5 . 
     The left and right input signals are processed collectively to provide a pair of spatially corrected left and right output signals. In particular, one embodiment equalizes the differences which exist between the two input signals in a manner which broadens and enhances the sound perceived by the listener. In addition, one embodiment adjusts the level of the sound which is common to both input signals so as to reduce clipping. Advantageously, one embodiment achieves sound enhancement with a simplified, low-cost, and easy-to-manufacture circuit which does not require separate circuits to process the common and differential signals as shown in  FIG. 5 . 
     Although some embodiments are described herein with reference to various sound enhancement system, the invention is not so limited, and can be used in a variety of other contexts in which it is desirable to adapt different embodiments of the sound enhancement system to different situations. To facilitate a complete understanding of the invention, the remainder of the detailed description is organized into the following sections and subsections: 
       FIG. 20  is a block diagram of a differential perspective correction apparatus  2002  from a first input signal  2010  and a second input signal  2012 . In one embodiment the first and second input signals  2010  and  2012  are stereo signals; however, the first and second input signals  2010  and  2012  need not be stereo signals and can include a wide range of audio signals. As explained in more detail below, the differential perspective correction apparatus  2002  modifies the audio sound information which is common to both the first and second input signals  2010  and  2012  in a different manner than the audio sound information which is not common to both the first and second input signals  2010  and  2012 . 
     The audio information which is common to both the first and second input signals  2010  and  2012  is referred to as the common-mode information, or the common-mode signal (not shown). In one embodiment, the common-mode signal does not exist as a discrete signal. Accordingly, the term common-mode signal is used throughout this detailed description to conceptually refer the audio information which exist in both the first and second input signals  2010  and  2012  at any instant in time. For example, if a one-volt signal is applied to both the first and second input signals  2010  and  2012 , the common-mode signal consists of one volt. 
     The adjustment of the common-mode signal is shown conceptually in the common-mode behavior block  2020 . The common-mode behavior block  2020  represents the alteration of the common-mode signal. One embodiment reduces the amplitude of the frequencies in the common-mode signal in order to reduce the clipping, which may result from high-amplitude input signals. 
     In contrast, the audio information which is not common to both the first and second input signals  2010  and  2012  is referred to as the differential information or the differential signal (not shown). In one embodiment, the differential signal is not a discrete signal, rather throughout this detailed description, the differential signal refers to the audio information which represents the difference between the first and second input signals  2010  and  2012 . For example, if the first input signal  2010  is zero volts and the second input signal  2012  is two volts, the differential signal is two volts (the difference between the two input signals  2010  and  2012 ). 
     The modification of the differential signal is shown conceptually in the differential-mode behavior block  2022 . As discussed in more detail below, the differential perspective correction apparatus  2002  equalizes selected frequency bands in the differential signal. That is, one embodiment equalizes the audio information in the differential signal in a different manner than the audio information in the common-mode signal. 
     The differential perspective correction apparatus  2002  spectrally shapes the differential signal in the differential-mode behavior block  2022  with a variety of filters to create an equalized differential signal. By equalizing selected frequency bands within the differential signal, the differential perspective correction apparatus  2002  widens a perceived sound image projected from a pair of loudspeakers placed in front of a listener. 
     Furthermore, while the common-mode behavior block  2020  and the differential-mode behavior block  2022  are represented conceptually as separate blocks, one embodiment performs these functions with a single, uniquely adapted system. Thus, one embodiment processes both the common-mode and differential audio information simultaneously. Advantageously, one embodiment does not require the complicated circuitry to separate the audio input signals into discrete common-mode and differential signals. In addition, one embodiment does not require a mixer which then recombines the processed common-mode signals and the processed differential signals to generate a set of enhanced output signals. 
     The differential perspective correction apparatus  2002  is in turn, connected to one or more output buffers  2006 . The output buffers  2006  output the enhanced first output signal  2030  and second output signal  2032 . As discussed in more detail below, the output buffers  2006  isolate the differential perspective correction apparatus  2002  from other components connected to the first and second output signals  2030  and  2032 . For example, the first and second output signals  2030  and  2032  can be directed to other audio devices such as a recording device, a power amplifier, a pair of loudspeakers and the like without altering the operation of the differential perspective correction apparatus  2002 . 
       FIG. 21  is a block diagram of a system that uses differential amplifiers to provide the differential perspective correction shown in  FIG. 20 . In  FIG. 21 , the first input  2010  is provided to a non-inverting input of a first differential amplifier  2102  and to a first input of a cross-over impedance block  2106 . The second input  2012  is provided to a non-inverting input of a second differential amplifier  2104  and to a second terminal of the cross-over impedance block  2106 . An inverting input of the first differential amplifier  2102  is provided to a first terminal of a cross-over impedance block  2107  and to a first terminal of a first feedback impedance  2108 . An output of the first differential amplifier  2102  is provided to the first output  2030  and to a second terminal of the first feedback impedance  2108 . An inverting input of the second differential amplifier  2104  is provided to a second terminal of the cross-over impedance block  2107  and to a first terminal of a second feedback impedance  2109 . An output of the second differential amplifier  2104  is provided to the second output  2032  and to a second terminal of the second feedback impedance  2109 . 
     The impedances of the blocks  2106 ,  2107 ,  2108  and  2109  are typically frequency dependent and may be constructed as filters using, for example, resistors, capacitors and/or inductors. In one embodiment, the impedances  2108  and  2109  are not frequency dependent. 
       FIG. 22  is an amplitude-versus-frequency chart, which illustrates the common-mode gain at both the left and right output terminals  2030  and  2032 . The common-mode gain is represented with a first common-mode gain curve  2200 . As shown in the common-mode gain curve  2200 , the frequencies below approximately 130 hertz (Hz) are de-emphasized more than the frequencies above approximately 130 Hz. For frequencies above approximately 130 Hz, the frequencies are uniformly reduced by approximately 6 decibels. 
       FIG. 23  illustrates the overall correction curve  2300  generated by the combination of the first and second cross-over networks  2106 , and  2107 . The approximate relative gain values of the various frequencies within the overall correction curve  2300  can be measured against a zero (0) dB reference. 
     With such a reference, the overall correction curve  2300  is defined by two turning points labeled as point A and point B. At point A, which in one embodiment is approximately 125 Hz, the slope of the correction curve changes from a positive value to a negative value. At point B, which in one embodiment is approximately 2 kHz, the slope of the correction curve changes from a negative value to a positive value. 
     Thus, the frequencies below approximately 125 Hz are de-emphasized relative to the frequencies near 125 Hz. In particular, below 125 Hz, the gain of the overall correction curve  2300  decreases at a rate of approximately 6 dB per octave. This de-emphasis of signal frequencies below 125 Hz prevents the over-emphasis of very low, (i.e. bass) frequencies. With many audio reproduction systems, over emphasizing audio signals in this low-frequency range relative to the higher frequencies can create an unpleasurable and unrealistic sound image having too much bass response. Furthermore, over emphasizing these frequencies may damage a variety of audio components including the loudspeakers. 
     Between point A and point B, the slope of one overall correction curve is negative. That is, the frequencies between approximately 125 Hz and approximately 2 kHz are de-emphasized relative to the frequencies near 125 Hz. Thus, the gain associated with the frequencies between point A and point B decrease at variable rates towards the maximum-equalization point of −8 dB at approximately 2 kHz. 
     Above 2 kHz the gain increases, at variable rates, up to approximately 20 kHz, i.e., approximately the highest frequency audible to the human ear. That is, the frequencies above approximately 2 kHz are emphasized relative to the frequencies near 2 kHz. Thus, the gain associated with the frequencies above point B increases at variable rates towards 20 kHz. 
     These relative gain and frequency values are merely design objectives and the actual figures will likely vary from system to system. Furthermore, the gain and frequency values may be varied based on the type of sound or upon user preferences without departing from the spirit of the invention. For example, varying the number of the cross-over networks and varying the resister and capacitor values within each cross-over network allows the overall perspective correction curve  2300  be tailored to the type of sound reproduced. 
     The selective equalization of the differential signal enhances ambient or reverberant sound effects present in the differential signal. As discussed above, the frequencies in the differential signal are readily perceived in a live sound stage at the appropriate level. Unfortunately, in the playback of a recorded performance the sound image does not provide the same 360 degree effect of a live performance. However, by equalizing the frequencies of the differential signal with the differential perspective correction apparatus  2002 , a projected sound image can be broadened significantly so as to reproduce the live performance experience with a pair of loudspeakers placed in front of the listener. 
     Equalization of the differential signal in accordance with the overall correction curve  2300  is intended to de-emphasize the signal components of statistically lower intensity relative to the higher-intensity signal components. The higher-intensity differential signal components of a typical audio signal are found in a mid-range of frequencies between approximately 2 to 4 kHz. In this range of frequencies, the human ear has a heightened sensitivity. Accordingly, the enhanced left and right output signals produce a much improved audio effect. 
     The number of cross-over networks and the components within the cross-over networks can be varied in other embodiments to simulate what are called head related transfer functions (HRTF). Head related transfer functions describe different signal equalizing techniques for adjusting the sound produced by a pair of loudspeakers so as to account for the time it takes for the sound to be perceived by the left and right ears. Advantageously, an immersive sound effect can be positioned by applying HRTF-based transfer functions to the differential signal so as to create a fully immersive positional sound field. 
     Examples of HRTF transfer functions which can be used to achieve a certain perceived azimuth are described in the article by E. A. B. Shaw entitled “Transformation of Sound Pressure Level From the Free Field to the Eardrum in the Horizontal Plane”, J. Acoust. Soc. Am., Vol. 56, No. 6, December 1974, and in the article by S. Mehrgardt and V. Mellert entitled “Transformation Characteristics of the External Human Ear”, J. Acoust. Soc. Am., Vol. 61, No. 6, June 1977. 
     Single Chip Implementation 
       FIG. 24  is a block diagram of one embodiment of a sound enhancement system  2400  that can be implemented on a single chip. As described in connection with  FIGS. 1-23  above, the system  2400  includes a vertical image enhancement block  2402 , a bass enhancement block  2404  and a horizontal image enhancement block  2406 . External connections to the system  2400  are provided through connector pins P 1 -P 27 . A positive supply voltage is provided to the pin P 25 , a negative supply voltage is provided to the pin P 26 , and a ground is provided to the pin P 27 . A first terminal of a compression coupling capacitor  2421  is provided to the pin P 10  and a second terminal of the compression coupling capacitor  2421  is provided to the pin P 11 . A first terminal of a compression delay capacitor  2420  is provided to the pin P 13  and a second terminal of the compression delay capacitor  2420  is provided to the pin P 14 . A first terminal of a width-control resistor  2430  is provided to the pin P 19  and a second terminal of the width-control resistor  2430  is provided to the pin P 20 . A first terminal of a width-control resistor  2431  is provided to the pin P 21  and a second terminal of the width-control resistor  2431  is provided to the pin P 22 . In one embodiment, the width-control resistors  2430  and  2431  are variable resistors. 
       FIG. 25A  is a schematic diagram of a left-channel of the vertical image enhancement block  2402 .  FIG. 25B  is a schematic diagram of a right-channel of the vertical image enhancement block  2402 . In  FIG. 25A , a left channel input is provided to the pin P 2  and left channel bypass input is provided to the pin P 1 . The pin P 1  is provided to a first terminal of a resistor  2501 . A second terminal of the resistor  2501  is provided to a first terminal of a resistor  2502  and to a first terminal of a capacitor  2503 . The pin P 2  is provided to a first terminal of a resistor  2504  and to a first terminal of a capacitor  2505 . A second terminal of the capacitor  2505  is provided to a first terminal of a resistor  2506  and to a first terminal of a resistor  2507 . A second terminal of the resistor  2506  is provided to ground. 
     A second terminal of the resistor  2502  is provided to a second terminal of the capacitor  2503 , to a second terminal of the resistor  2504 , to a second terminal of the resistor  2507  to a first terminal of a resistor  2508 , and to an inverting input of an operational amplifier (opamp)  2510 . A non-inverting input of the opamp  2510  is provided to ground. A second terminal of the resistor  2508  is provided to a first terminal of a resistor  2509  and to a first terminal of a capacitor  2512 . A second terminal of the resistor  2509  is provided to a second terminal of the capacitor  2512 , to an output of the opamp  2510 , and to a left-channel output  2511 . 
     In one embodiment, the resistor  2501  is 9.09 k ohms, the resistor  2502  is 27.4 k ohms, the capacitor  2503  is 0.1 uf, the resistor  2504  is 22.6 k ohms, the capacitor  2505  is 0.1 μf, the resistor  2506  is 3.01 k ohms, the resistor  2507  is 4.99 k ohms, the resistor  2508  is 9.09 k ohms, the resistor  2509  is 27.4 k ohms, the capacitor  2512  is 0.1 uf and the opamp  2510  is a TL074 type or equivalent. 
     The right-channel shown in  FIG. 25B  is similar to the left channel shown in  FIG. 25A , having a bypass input from the pin P 3 , a right-channel input from the pin P 4  and a right-channel output  2514 . 
       FIG. 26  is a schematic diagram of the bass enhancement block  2404 . The left-channel output  2511  from  FIG. 25A  is provided to a first terminal of a resistor  2601  and to a first terminal of a resistor  2611 . The right-channel output  2514  from  FIG. 25B  is provided to a first terminal of a resistor  2602  and to a first terminal of a resistor  2614 . 
     A second terminal of the resistor  2601  is provided to a second terminal of the resistor  2602 , to a first terminal of a resistor  2625 , and to a first terminal of a capacitor  2603 . A second terminal of the capacitor  2603  is provided to ground. A second terminal of the resistor  2625  is provided to an inverting input of an opamp  2606 , to a first terminal of a capacitor  2605  and to a first terminal of a resistor  2604 . A non-inverting input of the opamp  2606  is provided to ground. An output of the opamp  2606  is provided to a second terminal of the resistor  2604 , to a second terminal of the capacitor  2605 , and to an input of a filter block  2607  (shown in more detail in  FIG. 27 ). First, second, and third outputs of the filter block  2607  are provided to an inverting input of an opamp  2608  and to a first terminal of a resistor  2609 . A non-inverting input of the opamp  2608  is provided to ground. An output of the opamp  2608  is provided to a second terminal of the resistor  2609  and to the pin P 10 . 
     The pin P 10  is also provided to an input of a compressor  2610  (shown in more detail in  FIG. 28 ). An output of the compressor  2610  is provided to the pin P 12 . The pin P 12  is provided to the pin P 16 . The pin P 16  is provided to a first terminal of a resistor  2612  and to a first terminal of a resistor  2613 . 
     A second terminal of the resistor  2612  is provided to a second terminal of the resistor  2611 , to an inverting input of an opamp  2620  and to a first terminal of a resistor  2619 . A non-inverting input of the opamp  2620  is provided to ground. An output of the opamp  2620  is provided to a second terminal of the resistor  2619  and to a first terminal of the resistor  2621 . A second terminal of the resistor  2621  is provided to the pin P 17 . An output of the opamp  2620  is also provided as a left-channel output  2630 . 
     A second terminal of the resistor  2613  is provided to a second terminal of the resistor  2614 , to an inverting input of an opamp  2615  and to a first terminal of a resistor  2617 . A non-inverting input of the opamp  2615  is provided to ground. An output of the opamp  2615  is provided to a second terminal of the resistor  2617  and to a first terminal of the resistor  2618 . A second terminal of the resistor  2618  is provided to the pin P 18 . An output of the opamp  2615  is also provided as a right-channel output  2631 . 
     In one embodiment, the resistors  2601 ,  2602 , and  2604  are 43.2 k ohms, the capacitor  2603  is 0.022 uf, the resistor  2625  is 21.5 k ohms, and the capacitor  2605  is 0.01 uf. In one embodiment, the resistor  2609  is 100 k ohms, the resistors  2611 ,  2612 ,  2613 ,  2614 ,  2617 , and  2619  are 10 k ohms, and the resistors  2618  and  2621  are 200 ohms. In one embodiment, the opamps  2606 ,  2608 ,  2615 , and  2620  are TL074 types or equivalents thereof. 
       FIG. 27  is a schematic diagram of the filter system  2607 . In  FIG. 27 , the input is provided to a first terminal of resistors  2701 - 2704 . A second terminal of resistor  2701  is provided to a first terminal of a resistor  2710 , to a first terminal of a capacitor  2721  and to a first terminal of a capacitor  2720 . A second terminal of the capacitor  2721  is provided to a first terminal of a resistor  2722  and to an inverting input of an opamp  2732 . A non-inverting input of the opamp  2732  is provided to ground. An output of the opamp  2732  is provided to a second terminal of the capacitor  2720 , to a second terminal of the resistor  2722 , and to a first terminal of a resistor  2723 . A second terminal of the resistor  2723  is provided to the first filter output. 
     A second terminal of the resistor  2702  is provided to a first terminal of a resistor  2712  and to the pin P 5 . A second terminal of the resistor  2712  is provided to ground. 
     A second terminal of the resistor  2703  is provided to a first terminal of a resistor  2713  and to the pin P 7 . A second terminal of the resistor  2713  is provided to ground. 
     The pin P 6  is provided to a first terminal of a capacitor  2724  and to a first terminal of a capacitor  2728 . A second terminal of the capacitor  2728  is provided to a first terminal of a resistors  2725 , to a first terminal of a resistor  2726 , and to an inverting input of an opamp  2729 . A non-inverting input of the opamp  2729  is provided to ground. An output of the opamp  2729  is provided to a second terminal of the capacitor  2724 , to a second terminal of the resistor  2725 , to a second terminal of the resistor  2726 , and to a first terminal of a resistor  2730 . The second terminal of the capacitor  2724  is provided to the pin P 8 . A second terminal of the resistor  2725  is provided to the pin P 9 . A second terminal of the resistor  2730  is provided to the second filter output. 
     The second filter output is a low-frequency output (e.g., 40 Hz) when pin P 5  is shorted to pin P 6  and pins P 8  and P 9  are open. The second filter output is a high-frequency output (e.g., 150 Hz) when Pin P 7  is shorted to pin P 6  and pin P 8  is shorted to pin P 9 . 
     A second terminal of the resistor  2704  is provided to a first terminal of a resistor  2714 , to a first terminal of a capacitor  2731  and to a first terminal of a capacitor  2735 . A second terminal of the capacitor  2735  is provided to a first terminal of a resistor  2734  and to an inverting input of an opamp  2736 . A non-inverting input of the opamp  2736  is provided to ground. An output of the opamp  2736  is provided to a second terminal of the capacitor  2731 , to a second terminal of the resistor  2734  and to a first terminal of a resistor  2737 . A second terminal of the resistor  2737  is provided to the third filter output. 
     In one embodiment, the first filter output is a bandpass filter centered at 100 Hz, the third filter output is a bandpass filter centered at 60 Hz, and the second filter output is a bandpass filter centered at either 40 Hz or 150 Hz (as described above). 
     In one embodiment, the resistor  2701  is 31.6 k ohms, the resistor  2702  is 56.2 k ohms, the resistor  2703  is 21 k ohms, the resistor  2704  is 37.4 k ohms, the resistor  2710  is 4.53 k ohms, the resistor  2712  is 13 k ohms, the resistor  2713  is 3.09 k ohms, the resistor  2714  is 8.87 k ohms, the resistor  2722  is 63.4 k ohms, the resistor  2723  is 100 k ohms, the resistor  2725  is 57.6 k ohms, the resistor  2726  is 158 k ohms, the resistor  2730  is 100 k ohms, the resistor  2734  is 107 k ohms, and the resistor  2737  is 100 k ohms. In one embodiment, the capacitors  2720 ,  2721 ,  2724 ,  2728 ,  2731 , and  2735  are 0.1 uf. In one embodiment, the opamps  2732 ,  2729  and  2736  are TL074 types or equivalents thereof. 
       FIG. 28  is a schematic diagram of the compressor  2610 . The compressor  2610  includes a peak detector  2804 , a bias circuit  2802 , a gain control block  2806 , and an output buffer  2810 . The peak detector is built around a diode  2810  and a diode  2811 . The bias circuit is built around a transistor  2820  and a zener diode  2816 . The gain control circuit is built around a FET  2814 . The output buffer is built around an opamp  2824 . 
     The input to the compressor  2610  is provided at the pin P 10 . The pin P 10  is provided to a first terminal of a resistor  2827 . A second terminal of the resistor  2827  is provided to a drain of the FET  2814  and to a first terminal of a resistor  2822 . A second terminal of the resistor  2822  is provided to an inverting input of the opamp  2824  and to a first terminal of a resistor  2823 . A non-inverting input of the opamp  2824  is provided to ground. An output of the opamp  2824  is provided to a second terminal of the resistor  2823  and to the pin P 12 . The pin P 12  is the output of the compressor  2616 . 
     The source of the FET  2814  is provided to ground. The gate of the FET  2814  is provided to a first terminal of a resistor  2813 , to a first terminal of a resistor  2815 , and to the pin P 13 . The pin P 14  is provided to a second terminal of the resistor  2815 . 
     The second terminal of the resistor  2813  is provided to the cathode of the diode  2811 . The anode of the diode  2811  is provided to the cathode of the diode  2810  and to the pin P 11 . The anode of the diode  2810  is provided to a first terminal of a resistor  2812 . A second terminal of the resistor  2812  is provided to the pin P 14 . 
     The pin P 14  is also provided to a first terminal of a resistor  2818  and to the emitter of a PNP transistor  2820 . A second terminal of the resistor  2818  is provided to ground. The base of the PNP transistor  2820  is provided to a first terminal of a resistor  2817  and to a first terminal of a resistor  2819 . The second terminal of the resistor  2817  is provided to ground. The collector of the PNP transistor  2820  is provided to a second terminal of the resistor  2819 , to the anode of the zener diode  2816 , and to the pin P 15 . The cathode of the zener diode  2816  is provided to ground. The pin P 15  is provided to allow a current limiting bias resistor to be connected between the zener diode and the negative power supply voltage. 
     The capacitor  2421  connected between pin P 10  and P 11  AC coupling of the input to the peak detector circuit. The capacitor  2420  connected between pins P 13  and P 14  provides a delay time constant for the onset of compression. 
     In one embodiment, the diodes  2810  and  2811  are 1N4148 types or equivalent. In one embodiment, the FET  2814  is a 2N3819 or equivalent, the PNP transistor  2820  is a 2N2907 or equivalent, and the zener diode  2816  is a 3.3 volt zener (1N746A or equivalent). In one embodiment, the opamp  2824  is a TL074 type or equivalent. The capacitor  2420  is a DC block, and the capacitor  2421  sets the compression delay. In one embodiment, the resistor  2812  is 1 k ohms, the resistor  2813  is 10 k ohms, the resistor  2815  is 100 k ohms, the resistor  2817  is 4.12 k ohms, the resistor  2818  is 1.2 k ohms, the resistor  2819  is 806 ohms, the resistor  2822  is 10 k ohms, the resistor  2827  is 1 k ohms and the resistor  2823  is 100 k ohms. 
     The gain control block  2806  operates as a voltage controlled voltage divider. The voltage divider is formed by the resistor  2827  and the drain-to-source resistance of the FET  2814 . The drain-to-source resistance of the FET  2814  is controlled by the voltage applied to the gate of the FET  2814 . The output buffer  2810  amplifies the voltage produced by the voltage controlled voltage divider (that is, the voltage at the drain of the FET  2814 ) and provides an output voltage at the pin P 12 . The bias circuit  2802  biases the FET  2814  into a linear operating region. The peak detect circuit  2804  detects the peak magnitude of the signal provided at the pin P 10  and reduces the “gain” of the gain control  2806  (by changing the drain-to-source resistance of the FET  2814 ) in response to an increase in the peak magnitude. 
       FIG. 29  is a schematic diagram of the horizontal image enhancement block  2406 . In the block  2406 , the left-channel signal  2630  from the bass module  2404  is provided to a first terminal of a resistor  2903  and to a first terminal of a resistor  2901 . A second terminal of the resistor  2901  is provided to ground. The right-channel signal  2631  from the bass module  2404  is provided to a first terminal of a resistor  2904  and to a first terminal of a resistor  2902 . A second terminal of the resistor  2902  is provided to ground. 
     A second terminal of the resistor  2903  is provided to a first terminal of a resistor  2905  and to a non-inverting input of an opamp  2914 . A second terminal of the resistor  2904  is provided to a first terminal of a capacitor  2906  and to a non-inverting input of an opamp  2912 . A second terminal of the capacitor  2906  is provided to a second terminal of the resistor  2905 . 
     An inverting input of the opamp  2912  is provided to a first terminal of a capacitor  2911 , to a first terminal of a capacitor  2907 , to a first terminal of a capacitor  2910 , and to the pin P 21 . An output of the opamp  2912  is provided to a first terminal of a resistor  2913 , to the pin P 22 , and to a second terminal of the capacitor  2911 . 
     An inverting input of the opamp  2914  is provided to a first terminal of a capacitor  2915 , to the pin P 19 , to a first terminal of a resistor  2908 , and to a first terminal of a resistor  2909 . A second terminal of the resistor  2909  is provided to a second terminal of the capacitor  2910 . A second terminal of the resistor  2908  is provided to a second terminal of the capacitor  2907 . An output of the opamp  2914  is provided to a first terminal of a resistor  2917 , to the pin P 20 , and to a second terminal of the capacitor  2915 . 
     A second terminal of the resistor  2913  is provided to the pin P 24  as a right-channel output. A second terminal of the resistor  2917  is provided to the pin P 23  as a left-channel output. A variable resistor  2430  connected between the pins P 19  and P 20  controls the apparent spatial image width of the left channel. A variable resistor  2431  connected between the pins P 21  and P 22  controls the apparent spatial image width of the right channel. In one embodiment, the variable resistors  2930  and  2931  are mechanically connected such that varying one resistance also varies the other. 
     In one embodiment, the resistors  2901  and  2902  are 100 k ohms, the resistors  2903  and  2904  are 10 k ohms, the resistor  2905  is 8.66 k ohms, the resistor  2908  is 15 k ohms, the resistor  2909  is 30.1 k ohms, and the resistors  2917  and  2913  are 200 ohms. In one embodiment, the capacitor  2906  is 0.018 uf, the capacitor  2907  is 0.001 uf, the capacitor  2910  is 0.082 uf and the capacitors  2915  and  2911  are 22 pf. In one embodiment, the variable resistors  2430  and  2431  have a maximum resistance of 100 k ohms. In one embodiment, the opamps are TL074 types or equivalent. 
       FIG. 30  is a schematic diagram of a correction system  3000 , which can be used as the stereo image enhancement system  124 . The system  3000  includes a differential amplifier, which provides a common-mode behavior  3020  and a differential-mode behavior  3022 . 
     The system  3000  includes two transistors  3010  and  3012 ; multiple capacitors  3020 ,  3022 ,  3024 ,  3026  and  3028 ; and multiple resistors  3040 ,  3042 ,  3044 ,  3046 ,  3048 ,  3050 ,  3052 ,  3054 ,  3056 ,  3058 ,  3060 ,  3062  and  3064 . Located between the transistors  3010  and  3012  are three crossover networks  3070 ,  3072  and  3074 . The first crossover network  3070  includes the resistor  3060  and the capacitor  3024 . The second crossover network  3072  includes the resistor  3062  and the capacitor  3026 , and the third crossover network  3074  includes the resistor  3064  and the capacitor  3028 . 
     A left input terminal  3000  (LEFT IN) provides a left input signal to the base of transistor  3010  through the capacitor  3020  and the resistor  3040 . A power supply V CC    3040  is connected to the base of transistor  3010  through the resistor  3046 . The power supply V CC    3040  is also connected to the collector of transistor  3010  through the resistor  3046 . The base of the transistor  3010  is also connected to a ground  3041  through the resistor  3044  while the emitter of transistor  3010  is connected to the ground  3041  through the resistor  3048 . 
     The capacitor  3020  is a decoupling capacitor that provides direct current (DC) isolation of the input signal at the left input terminal  3000 . The resistors  3042 ,  3044 ,  3046  and  3048 , on the other hand, create a bias circuit that provides stable operation of the transistor  3010 . In particular, the resistors  3042  and  3044  set the base voltage of transistor  3010 . The resistor  3046  in combination with the third crossover network  3074  together set the DC value of the collector-to-emitter voltage of the transistor  3010 . The resistor  3048  in combination with the first and second crossover networks  3070  and  3072  together set the DC current of the emitter of the transistor  3010 . 
     In one embodiment, the transistor  3010  is an NPN 2N2222A transistor which is commonly available from a wide variety of transistor manufacturers. The capacitor  3020  is 0.22 microfarads. The resistors  3040  is 22 kilohms (kohm), the resistor  3042  is 41.2 kohm, the resistor  3046  is 10 kohm, and the resistor  3048  is 6.8 kohm. One of ordinary skill in the art will recognize, however, that a variety of transistors, capacitors and resistors with different values can be used. 
     The right input terminal  3002  provides a right input signal to the base of the transistor  3012  through the capacitor  3022  and the resistor  3050 . The power supply V CC    3040  is connected to the base of transistor  3012  through the resistor  3052 . The power supply V CC    3040  is also connected to the collector of transistor  3012  through the resistor  3056 . The base of the transistor  3012  is also connected to the ground  3041  through the resistor  3054  while the emitter of the transistor  3012  is connected to the ground  3041  through the resistor  3058 . 
     The capacitor  3022  is a decoupling capacitor that provides direct current (DC) isolation of the input signal at the right input terminal  3002 . The resistors  3052 ,  3054 ,  3056  and  3058 , on the other hand, create a bias circuit that provides stable operation of the transistor  3012 . In particular, the resistors  3052  and  3054  set the base voltage of transistor  3012 . The resistor  3056  in combination with the third crossover network  3074  together set the DC value of the collector-to-emitter voltage of the transistor  3012 . The resistor  3058  in combination with the first and second crossover networks  3070  and  3072  together set the DC current of the emitter of the transistor  3012 . 
     In one embodiment, the transistor  3012  is an NPN 2N2222A transistor which is commonly available from a wide variety of transistor manufacturers. The capacitor  3022  is 0.22 microfarads. The resistors  3050  is 22 kilohms (kohm), the resistor  3052  is 41.2 kohm, the resistor  3056  is 10 kohm, and the resistor  3058  is 6.8 kohm. One of ordinary skill in the art will recognize however, that a variety of transistors, capacitors and resistors with different values can be used. 
     The system  3000  creates two types of voltage gains, a common-mode voltage gain and a differential voltage gain. The common-mode voltage gain is a change in the voltage that is common to both the left and right input terminals  3000  and  3002 . The differential gain is a change in the output voltage due to the difference between the voltages applied to the left and right input terminals  3000  and  3002 . 
     In the system  3000 , the common-mode gain is designed to reduce clipping that may result from high-amplitude input signals. In one embodiment, the common-mode gain at the left output terminal  3004  is primarily defined by the resistors  3040 ,  3042 ,  3044 ,  3046  and  3048 . In one embodiment, the common-mode gain is approximately six decibels. 
     The frequencies below approximately 30 hertz (Hz) are de-emphasized more than the frequencies above approximately 30 Hz. For frequencies above approximately 30 Hz, the frequencies are uniformly reduced by approximately 6 decibels. 
     The common-mode gain, however, may vary for or a given implementation by varying the values of the resistors  3040 ,  3042 ,  3044 ,  3050 ,  3052  and  3054 . 
     The differential gain between the left and right output terminals  3004  and  3006  is defined primarily by the ratio of the resistors  3046  and  3048 , the ratio of the resistors  3056  and  3058 , and the three crossover networks  3070 ,  3072  and  3074 . As discussed in more detail below, one embodiment equalizes certain frequency ranges in the differential input. Thus, the differential gain varies based on the frequency of the left and right input signals. 
     Because the crossover networks  3070 ,  3072  and  3074  equalize the frequency ranges in the differential input, the frequencies in the differential signal can be altered without affecting the frequencies in the common-mode signal. As a result, one embodiment can create enhanced audio sound in an entirely unique and novel manner. Furthermore, the differential perspective correction apparatus  102  is much simpler and cost-effective to implement than many other audio enhancement systems. 
     Focusing now on the three crossover networks  3070 ,  3072  and  3074 , the crossover networks  3070 ,  3072  and  3074  act as filters which spectrally shape the differential signal. A filter is usually characterized as having a cut-off frequency, which separates a passband of frequencies from a stopband of frequencies. The cut-off frequency is the frequency, which marks the edge of the passband and the beginning of the transition to the stopband. Typically, the cut-off frequency is the frequency, which is de-emphasized by three decibels relative to other frequencies in the passband. The passband of frequencies are those frequencies which pass through a filter with essentially no equalization or attenuation. The stopband of frequencies, on the other hand, are those frequencies, which the filter equalizes or attenuates. 
       FIG. 31  shows one embodiment of the present invention with just the first crossover network  3070 . The first crossover network  3070  comprises the resistor  3060  and the capacitor  3024 , which interconnect the emitters of transistors  3010  and  3012 . Because the first crossover network  3070  equalizes frequencies in the lower portion of the frequency spectrum, it is thus called a high-pass filter. In one embodiment, the value of the resistor  3060  is approximately 27.01 kohm and the value of the capacitor  3024  is approximately 0.68 microfarads. 
     The values of the resistor  3060  and the capacitor  3024  are selected to define a cut-off frequency in a low range of frequencies. In one embodiment, the cut-off frequency is approximately 78 Hz, a stopband below approximately 78 Hz and a passband above approximately 78 Hz. Frequencies below approximately 78 Hz are de-emphasized relative to frequencies above approximately 78 Hz. However, because the first crossover network  3070  is only a first-order filter, frequencies defining the cut-off frequency are design goals. The exact characteristic frequencies may vary for a given implementation. Furthermore, other values for the resistor  3060  and the capacitor  3024  can be chosen to vary the cut-off frequency in order to de-emphasize other desired frequencies. 
       FIG. 32  is a schematic diagram of a differential perspective correction apparatus  3200  with both the second and third crossover networks  3070  and  3072 . Like the first crossover network  3070 , the second crossover network  3072  is also preferably a filter, which equalizes certain frequencies in the differential signal. Unlike the first crossover network  3070 , however, the second crossover network  3072  is a high-pass filter which also de-emphasizes lower frequencies in the differential signal relative to the higher frequencies in the differential signal. 
     As shown in  FIG. 32 , the second crossover network  3072  interconnects the emitters of transistors  3010  and  3012 . In addition, the second crossover network  3072  comprises the resistor  3062  and the capacitor  3026 . Preferably, the value of the resistor  3062  is approximately 1 kohm and the value of the capacitor  3026  is approximately 0.01 microfarads. 
     These values are selected to define a cut-off frequency in a high range of frequencies. In one embodiment, the cut-off frequency is approximately 15.9 kilohertz (kHz). Frequencies in the stopband below approximately 15.9 kHz are de-emphasized relative to frequencies in the passband above 15.9 kHz. 
     However, because the second crossover network  3072 , like the first crossover network  3070 , is a first-order filter, frequencies defining the passband are design goals. The exact characteristic frequencies may vary for a given implementation. Furthermore, other values for the resistor  3062  and capacitor  3026  can be chosen to vary the cut-off frequency so as to de-emphasize other desired frequencies. 
     Referring now to  FIG. 33 , the third crossover network  3074  interconnects the collectors of transistors  3010  and  3012 . The third crossover network  3074  includes the resistor  3064  and the capacitor  3028  which are selected to create a low-pass filter which de-emphasizes frequencies above a mid-range of frequencies. In one embodiment, the cut-off frequency of the low-pass filter is approximately 795 Hz. Preferably, the value of resistor  3064  is approximately 9.09 kohm and the value of the capacitor  3028  is approximately 0.022 microfarads. 
     In the correction generated by the third crossover network  3074  frequencies in the stopband above approximately 795 Hz are de-emphasized relative to frequencies in the passband below approximately 795 Hz. As discussed above, because the third crossover network  3074  is only a first-order filter, frequencies defining the low-pass filter in the third crossover network  3074  are design goals. The frequencies may vary for or given implementation. Furthermore, other values for resistor  3064  and capacitor  3028  can be chosen to vary the cut-off frequency so as to de-emphasize other desired frequencies. 
     In operation, the first, second and third crossover networks  3070 ,  3072  and  3074  work in combination to spectrally shape the differential signal. 
     The overall correction curve  2300  (shown in  FIG. 23 ) is defined by two turning points labeled as point A and point B. At point A, which in one embodiment is approximately 125 Hz, the slope of the correction curve changes from a positive value to a negative value. At point B, which in one embodiment is approximately 1.8 kHz, the slope of the correction curve changes from a negative value to a positive value. 
     Thus, the frequencies below approximately 125 Hz are de-emphasized relative to the frequencies near 125 Hz. In particular, below 125 Hz, the gain of the overall correction curve  2300  decreases at a rate of approximately 6 dB per octave. This de-emphasis of signal frequencies below 125 Hz prevents the over-emphasis of very low, (i.e., bass) frequencies. With many audio reproduction systems, over emphasizing audio signals in this low-frequency range relative to the higher frequencies can create an unpleasurable and unrealistic sound image having too much bass response. Furthermore, over emphasizing these frequencies may damage a variety of audio components, including the loudspeakers. 
     Between point A and point B, the slope of one overall correction curve is negative. That is, the frequencies between approximately 125 Hz and approximately 1.8 kHz are de-emphasized relative to the frequencies near 125 Hz. Thus, the gain associated with the frequencies between point A and point B decrease at variable rates towards the maximum-equalization point of −8 dB at approximately 1.8 kHz. 
     Above 1.8 kHz the gain increases, at variable rates, up to approximately 20 kHz, i.e., approximately the highest frequency audible to the human ear. That is, the frequencies above approximately 1.8 kHz are emphasized relative to the frequencies near 1.8 kHz. Thus, the gain associated with the frequencies above point B increases at variable rates towards 20 kHz. 
     These relative gain and frequency values are merely design objectives and the actual figures will likely vary from circuit to circuit depending on the actual value of components used. Furthermore, the gain and frequency values may be varied based on the type of sound or upon user preferences without departing from the spirit of the invention. For example, varying the number of the crossover networks and varying the resistor and capacitor values within each crossover network allows the overall perspective correction curve  2300  be tailored to the type of sound reproduced. 
     The selective equalization of the differential signal enhances ambient or reverberant sound effects present in the differential signal. As discussed above, the frequencies in the differential signal are readily perceived in a live sound stage at the appropriate level. Unfortunately, in the playback of a recorded performance the sound image does not provide the same 360-degree effect of a live performance. However, by equalizing the frequencies of the differential signal, a projected sound image can be broadened significantly so as to reproduce the live performance experience with a pair of loudspeakers placed in front of the listener. 
     Equalization of the differential signal in accordance with the overall correction curve  2300  is intended to de-emphasize the signal components of statistically lower intensity relative to the higher-intensity signal components. The higher-intensity differential signal components of a typical audio signal are found in a mid-range of frequencies between approximately 1 to 4 kHz. In this range of frequencies, the human ear has a heightened sensitivity. Accordingly, the enhanced left and right output signals produce a much-improved audio effect. 
     The number of crossover networks and the components within the crossover networks can be varied in other embodiments to simulate head related transfer functions (HRTF). Advantageously, an immersive sound effect can be positioned by applying HRTF-based transfer functions to the differential signal so as to create a fully immersive positional sound field. 
       FIG. 33  shows a differential perspective correction apparatus  3300  that allows a user to vary the amount of overall differential gain. In this embodiment, a fourth crossover network  3301  interconnects the emitters of transistors  3010  and  3012 . In this embodiment, the fourth crossover network  3301  comprises a variable resistor  3302 . 
     The variable resistor  3302  acts as a level-adjusting device and is ideally a potentiometer or similar variable-resistance device. Varying the resistance of the variable resistor  3302  raises and lowers the relative equalization of the overall perspective correction circuit. Adjustment of the variable resistor is typically performed manually so that a user can tailor the level and aspect of the differential gain according to the type of sound reproduced, and based on the user&#39;s personal preferences. Typically, a decrease in the overall level of the differential signal reduces the ambient sound information creating the perception of a narrower sound image. 
       FIG. 34  illustrates a differential perspective correction apparatus  3400  that allows a user to vary the amount of common-mode gain. The differential perspective correction apparatus  3400  includes contains a fourth crossover network  3401 . The fourth crossover network  3401  includes a resistor  3402 , a resistor  3404 , a capacitor  3406  and a variable resistor  3408 . The capacitor  3406  removes the differential information and allows the variable resistor and resisters  3402  and  3404  to vary the common-mode gain. 
     The resisters  3402  and  3404  can be a wide variety of values depending on the desired range of common-mode gain. The variable resistor  3408 , on the other hand, acts as a level-adjusting device, which adjusts the common-mode gain within the desired range. Ideally, the variable resistor  3408  is a potentiometer or similar variable-resistance device. Varying the resistance of the variable resistor  3408  affects both transistors  3010  and  3012  equally and thereby raises and lowers the relative equalization of the overall common-mode gain. 
     Adjustment of the variable resistor is typically performed manually so that a user can tailor the level and aspect of the common-mode gain. An increase in the common-mode gain emphasizes the audio information, which is common to both input signals  3002  and  3004 . For example, increasing the common-mode gain in a sound system will emphasize the audio information at the center stage positioned between a pair of loudspeakers. 
       FIG. 35  illustrates a differential perspective correction apparatus  3500  that has a first crossover network  3501  located between the emitters of transistors  3010  and  3012  and a second crossover network  3502  located between the collectors of transistors  3010  and  3012 . 
     The first crossover network  3501  is a high-pass filter which de-emphasizes frequencies in the lower portion of the frequency spectrum. In this embodiment, the first crossover network  3501  comprises a resistor  3510  and a capacitor  3512 . The values of the resistor  3510  and the capacitor  3512  are selected to define a high-pass filter with a cut-off frequency of approximately 350 Hz. Accordingly, the value of resistor  3510  is approximately 27.01 kohm and the value of the capacitor  3512  is approximately 0.15 microfarads. In operation, the frequencies below 30 Hz are de-emphasized relative to the frequencies above 350 Hz. 
     The second crossover network  3502  interconnects the collectors of transistors  3010  and  3012 . The second crossover network  3502  is a low-pass filter which de-emphasizes frequencies in the lower portion of the frequency spectrum. In this embodiment, the second crossover network  3502  comprises a resistor  3520  and a capacitor  3522 . 
     The values of the resistor  3520  and the capacitor  3522  are selected to define a low-pass filter with a cut-off frequency of approximately 27.3 kHz. Accordingly, the value of the resistor  3520  is approximately 9.09 kohm and the value of the capacitor  3522  is approximately 0.0075 microfarads. In operation, the frequencies above 27.3 kHz are de-emphasized relative to the frequencies below 27.3 kHz. 
     The first and second crossover networks  3501  and  3502  work in combination to spectrally shape the differential signal. The frequencies below approximately 5 kHz are de-emphasized relative to the frequencies near 5 kHz. In particular, below 5 kHz, the gain of the overall correction increases at a rate of approximately 5 dB per octave. Furthermore, above 5 kHz, the gain of the overall correction curve  1400  also decreases at a rate of approximately 5 dB per octave. 
     The above embodiments of a differential perspective correction apparatus can also include output buffers  3630  as illustrated in  FIG. 36 . The output buffers  3630  are designed to isolate the perspective correction differential apparatus from variations in the load presented by a circuit connected to the left and right output terminals  3004  and  3006 . For example, when the left and right output terminals  3004  and  3006  are connected to a pair of loudspeakers, the impedance load of the loudspeakers will not alter the manner in which the differential perspective correction apparatus equalizes the differential signal. Accordingly, without the output buffers  3630 , circuits, loudspeakers and other components will affect the manner in which the differential perspective correction apparatus  102  equalizes the differential signal. 
     In one embodiment, the left output buffer  3630 A includes a left output transistor  3601 , a resistor  3604  and a capacitor  3604 . The power supply V CC    3040  is connected directly to the collector of transistor  3601 . The collector of transistor  3601  is connected to the ground  3041  through the resistor  3604  and to the left output terminal  3004  through the capacitor  3602 . In addition, the base of transistor  3601  is connected to the collector of transistor  3010 . 
     In one embodiment, the transistor  3601  is an NPN 2N2222A transistor, the resistor  3604  is 1 kohms and the capacitor  3602  is 0.22 microfarads. The resistor  3604 , the capacitor  3602  and the transistor  3601  create a unity gain. That is, the left output buffer  3630 A primarily passes the enhanced sound signals to the left output terminal  3004  without further equalizing the enhanced sound signals. 
     Likewise, one right output buffer  3630 B includes a right output transistor  3610 , a resistor  3612  and a capacitor  3614 . The power supply V CC    3040  is connected directly to the collector of the transistor  3610 . The collector of transistor  3610  is connected to the ground  3041  through the resistor  3612  and to the right output terminal through the capacitor  3614 . In addition, the base of transistor  3610  is connected to the collector of transistor  3012 . 
     In one embodiment, the transistor  3610  is an NPN 2N2222A transistor, the resistor  3612  is 1 kohm and the capacitor  3614  is 0.22 microfarads. The resistor  3612 , the capacitor  3614  and the transistor  3610  create a unity gain. That is, the right output buffer  3630 B primarily passes the enhanced sound signals to the right output terminal  3006  without further equalizing the enhanced sound signals. 
     One skilled in the art will recognize that the output buffers  3630  can also be implemented using other amplifiers, such as, for example, opamps and the like. 
       FIG. 37  shows yet another embodiment of the stereo image enhancement processor  124 . In  FIG. 37 , the left input  2630  is provided to a first terminal of a resistor  3710 , to a first terminal of a resistor  3716 , and to a first terminal of a resistor  3740 . The second terminal of the resistor  3710  is provided to a first terminal of a resistor  3711 , and to an inverting input of an opamp  3712 . The right input  2631  is provided to a first terminal of a resistor  3713 , to a first terminal of a resistor  3741 , and to a first terminal of a resistor  3746 . The second terminal of the resistor  3713  is provided to a first terminal of the resistor  3714  and to a non-inverting input of the opamp  3712 . The second terminal of the resistor  3714  is provided to ground. The second terminal of the resistor  3740  and a second terminal of the resistor  3741  are provided to a non-inverting input of the opamp  3744 , and to a first terminal of the resistor  3742 . The second terminal of the resistor  3742  provided to ground. 
     The output of the opamp  3744  in provided a first terminal of the resistor  3761 . A second terminal of the resistor  3761  is provided to an inverting input of the opamp  3744 . The second terminal of the resistor  3743  is provided to ground. Returning to the opamp  3712 , an output of the opamp  3712  is provided to a second terminal of the resistor  3711 . The output of the opamp  3712  is also provided in first terminal of the resistor  3715 . The second terminal of the resistor  3715  provided to a first terminal of a capacitor  3717 . A second terminal of the capacitor  3717  is provided to a first terminal of the resistor  3718 , to a first terminal of the resistor  3719 , to a first terminal of a capacitor  3721 , and to a first terminal of a resistor  3722 . The second terminal of the resistor  3718  is provided to ground. The second terminal of the resistor  3719  is provided to a second terminal of the resistor  3720 , and to the second terminal of the resistor  3725 . The second terminal of the capacitor  3721  is provided to a first terminal of the resistor  3720  and to a first terminal of the resistor  3023 . The second terminal of the resistor  3722  is provided to a first terminal of the resistor  3725  and to a first terminal of a capacitor  3724 . The second terminal of the resistor  3023  and the second terminal of the capacitor  3024  are both provided to ground. 
     The second terminal of the resistor  3719  is also provided to a first terminal of a resistor  3726  and to an inverting input of an opamp  3727 . A non-inverting input of the opamp  3727  is provided to ground. The second terminal of the resistor  3726  is provided to an output of the opamp  3727 . The output of the opamp  3727  is provided to a first fixed terminal of a potentiometer  3728 . A second fixed terminal of the potentiometer  3728  is provided ground. A wiper of the potentiometer  3728  is provided to the second terminal of a resistor  3747  and to a first terminal of a resistor  3729 . 
     An output of the opamp  3744  is provided to a first fixed terminal of a potentiometer  3745 . A second fixed terminal of the potentiometer  3745  is provided to ground. A wiper of the potentiometer  3745  is provided to the first terminal of the resistor  3730  and to a first terminal of the resistor  3751 . A second terminal of the resistor  3747  is provided to a first terminal of a resistor  3748  and to an inverting input of an opamp  3749 . 
     A non-inverting input of the opamp is  3749  provided to ground. An output of the opamp  3749  is provided to second terminal of the resistor  3748  and to the first terminal of the resistor  3750 . The second terminal of the resistor  3750  is provided to a second terminal of the resistor  3729 . A second terminal of the resistor  3730  provided to a non-inverting input of the opamp  3753 . A first terminal of the resistor  3731  is also provided to the non-inverting input of the opamp  3735 . The second terminal of the resistor  3731  is provided to ground. An inverting input of the opamp  3735  is provided to a first terminal of a resistor  3734  and to a first terminal of a resistor  3732 . The second terminal of the resistor  3732  provided to ground. An output of the opamp  3735  provided to a second terminal of a resistor  3734 . A second terminal of the resistor  3750 , a second terminal of the resistor  3751 , a second terminal of the resistor  3746 , and a first terminal of a resistor  3752  are all provided to a non-inverting input of an opamp  3755 . A second terminal of the resistor  3752  is provided to ground. A non-inverted input of the opamp  3755  is provided to a first terminal of a resistor  3753  and to a first terminal of a resistor  3754 . An output of the opamp  3755  is provided to a second terminal of the resistor  3754 . 
     The output of the opamp  3735  is provided as a left channel output and the output of the opamp  3755  is provided as a right channel output. 
     The resistors  3710 ,  3711 ,  3713 ,  3714 ,  3740 ,  3741 ,  3742 ,  3743 ,  37  and  3761  are all 33.2 K ohm resistors. The resistors  3716  and  3746  are both 80.6 K ohms. The potentiometers  3745  and  3728  are both 10.0 K linear potentiometers. The resistor  3715  is 1.0 K, the capacitor  3717  is 0.47 uf, the resistor  3718  is 4.42 K, the resistor  3719  is 121 K, the capacitor  3721  is 0.0047 uf, the resistor  3720  is 47.5 K, the resistor  3722  is 1.5 K, the resistor  3723  is 3.74 K, the resistor  3725  is 33.2 K., and the capacitor  3724  is 0.47 uf. The resistor  3726  is a 121 K. The resistors  3747  and  3748  are both 16.2 K. The resistors  3729  and  3750  are both 11.5 K. The resistors  3730  and  3751  are both 37.9 K. The resistors  3731 ,  3732 ,  3752 , and  3753 , are all 16.2 K. The resistor  3734  and  3754  are both 38.3 K. The opamps  3712 ,  3744 ,  3727 ,  3749 ,  3735 , and  3755  are all TL074 types or equivalents. 
     Digital Signal Processor Implementation 
     The acoustic correction system can also be readily implemented in software as described in connection with  FIG. 3 . Suitable processors include general purpose processors, Digital Signal Processors (DSP), and the like. 
       FIG. 38  is a block diagram of a software embodiment of the acoustic correction system  120 . In  FIG. 38 , a left-channel input  3801  is provided in input of a 10 db attenuator  3803 . An output of the attenuator  3803  is provided to an input of a filter  3804  and to a first throw of a DPDT switch  3805 . An output of the filter  3804  is provided to a second throw of the switch  3805 . A right-channel input  3802  is provided to an input of a 10 db attenuator  3806 . An output of the attenuator  3806  provided to an input of a filter  3807 , and to a first throw of the switch  3805 . An output of the filter  3807  is provided a second throw of the switch  3805 . 
     A first pole of the switch  3805  is provided to a first input of a summer  3828  and to a first input of a summer  3808 . A second poll of the switch  3805  is provided to a first input of a summer  3829  and to a second input of the summer  3808 . An output of the summer  3808  is provided to an input of the low pass filter  3809 . An output of the low pass filter  3809  is provided to an input of a dual-band bandpass filter  3810 , to an input of a dual-band bandpass filter  3811  and to an input of a 100 Hz band pass filter  3812 . 
     An output of the filter  3810  is provided to a first input of a summer  3821 , an output of the filter  3811  is provided the second input of the summer  3821 , and an output of the filter  3812  provided to a third input of the summer  3812 . An output of the summer  3821  is provided to an input of a 2.75 dB amplifier  3863 , to a first input of a multiplier  3824 , and to an input of an absolute-value block  3822 . An output of the absolute-value block  3822  is provided in input of a Fast Attack Slow Decay (FASD) compressor  3823 . An output of the FASD compressor  3823  is provided to a second input of the multiplier  3824 . 
     An output of the amplifier  3863  is provided to a positive input of a subtractor  3825 . An output of the multiplier  3824  provided to a negative input of the subtractor  3825 . An output of the subtractor  3825  is provided to a first input of a multiplier  3826 . An output of a bass control  3827  is provided to second input of the multiplier  3826 . An output of the multiplier  3826  is provided through a SPDT switch  3860  to a second input of the summer  3828  and to a second input of the summer  3829 . 
     An output of the summer  3828  is provided to a first input of a summer  3830 , to an input of a 9 dB attenuator  3833 , to a positive input of a subtractor  3837 , and to a first throw of a DPDT switch  3836 . An output of the summer  3829  is provided to a negative input of the subtractor  3837 , to a second input of the summer  3830 , to a input of a 9 db attenuator  3834 , and to a first throw of the switch  3836 . 
     An output of the summer  3830  is provided to an input of a 5 dB attenuator  3832 . An output the attenuator  3832  provided to first input of a summer  3835  and to a first input of a summer  3866 . An output of the attenuator  3833  is provided to a second input of the summer  3835 . An output of the attenuator  3834  is provided to a second input of the summer  3866 . An output of the summer  3835  provided to a second throw of the switch  3836 . An output of the summer  3866  is provided to a second throw of the switch  3836 . 
     An output of this subtractor  3837  is provided to an input of a 48 Hz highpass filter  3838 . An output of the high pass filter  3838  is provided to an input of a 6 dB attenuator  3840 , to an input of a 7 kHz highpass filter  3841 , and to an input of a 200 Hz lowpass filter  3842 . An output of the attenuator  3840  is provided the first input of a summer  3844 , an output of the highpass filter  3841  is provided to a second input of the summer  3844 , and an output of the low pass filter  3842  is provided through a 3 db attenuator  3843  to a third input of the summer  3844 . An output of the summer  3844  is provided to a first input of a multiplier  3845 . An output of a width control  3846  is provided to a second input of the multiplier  3845 . An output of the multiplier  3845  is provided to a third input of the summer  3835 , and through an inverter (i.e., a gain of −1) to a third input of the summer  3866 . 
     The first pole of the switch  3836  provided to a left channel output  3850 . A second pole of the switch  3836  is provided to a right output  3851 . 
     As shown in  FIG. 38 , left and right stereo input signals are provided to left and right inputs  3801  and  3802  respectively. For the bass enhancement portion of the processing (corresponding to the bass enhancement block  101  shown in  FIG. 1 ), the left and right channels are added together by the summer  3808 , processed as a monophonic signal, then added back into left and right channels by the summers  3828  and  3829  to form an enhanced stereo signal. The bass information is processed as a monophonic signal because there is typically little stereo separation in a bass frequency signal, so there is little need to duplicate the processing for the two channels. 
       FIG. 38  shows software user controls including: a software control  3827  to control the amount of bass enhancement, a software control  3846  to control the width of the apparent sound stage, as well as software switches  3805 ,  3860 , and  3836  to individually enable or disable the vertical, bass, and width image enhancements respectively. Depending on the application, these user controls can be either dynamically changeable or fixed to a specific configuration. The user controls can be “connected” to controls such as sliders, check boxes, and the like, in dialog box to allow the user to control the operation of the acoustic correction system. 
     In  FIG. 38 , the left and right inputs  3801  and  3802  are first processed with a gain of −10 dB to set the bypass level and prevent the signal from saturating during the processing that follows. Each channel is then processed through an elevation filter (filters  3804  and  3807  for left and right respectively) that performs the soundstage elevation and expansion as described in connection with  FIGS. 4-6 . 
     After the elevation filters, the left and right channels are mixed together and routed through the low pass filter  3809  followed by the bank of bandpass filters  3810 - 3812 . The low pass filter  3809  has a cutoff frequency of 284 Hz. Each of the following three filters  3810 - 3812  is a second order band pass filter. The filter  3810  is selectable as either 40 Hz or 150 Hz. The filter  3811  is selectable as either 60 Hz or 200 Hz. Thus, there are three useful configurations for speaker size: small, medium and large. All three configurations use the three band pass filters, but with different center frequencies for the filters  3810  and  3811 . 
     The outputs of the three active filters are then summed together by the summer  3821  and the sum is provided to the bass control stage. 
     The bass control stage includes an expander circuit having the absolute value detector  3822 , the fast attack slow decay peak detector  3823  and the multiplier  3824 . The output of the peak detector  3823  is used as a multiplier for the expander input signal to expand the dynamic range of the signal. 
     The second part of the bass control stage subtracts an expanded version of the stage&#39;s input signal from the same input signal with a 2.75 dB gain applied by the amplifier  3863 . This has the effect of limiting the level of high amplitude signals while adding a small constant gain to lower amplitude signals. 
     The output of the bass control stage is added into both the left channel signal and the right channel signal by the summers  3828  and  3829  respectively. The amount of enhanced bass signal that is mixed into the left and right channels is determined by the Bass Control  3827 . 
     The resulting left and right channel signals are then summed together by the summer  3830  to form a L+R signal, and subtracted by the subtractor  3837  to form a L−R signal. The L−R signal is shaped spectrally by processing it through the perspective curve (see  FIG. 7 ), which is implemented with a network of filters and gain adjustments as follows. First, the signal passes through the 48 Hz high pass filter  3838 . The output of this filter is then split and passed through the 7 kHz high pass filter  3841  and the 200 Hz low pass filter  3842 . Then the three filter outputs are summed together by the summer  3844  to form the perspective curve signal, using the following gain adjustments: −6 dB for the 48 Hz high pass filter  3838 , 0 dB (no adjustment) for the 7 kHz high pass filter  3841  and +3 dB for the 200 Hz low pass filter  3842 . The Width Control  3846  determines the amount of perspective curve signal that is passed through to the final summers  3835  and  3866 . 
     Finally, the left channel, right channel, L+R and L−R signals are mixed together by the summers  3835  and  3866  to produce the final left and right channel outputs respectively. The left channel output is formed by mixing the L+R signal with a −5 dB gain adjustment, the left channel signal with a −9 dB gain adjustment, and the perspective curve signal with no gain adjustment other than the gain adjustment provided by the Width Control  3846 . The right channel output is formed by mixing the L+R signal with a −5 dB gain adjustment, the right channel with a −9 dB gain adjustment, and an inverted perspective curve signal with no gain adjustment other than the Width Control. 
     The algorithm for the Fast Attack Slow Decay (FASD) Peak Detector  3823  is represented in pseudocode as follows: 
                                                if [in &gt; out(previous)] then                out = in − [[in − out(previous)] * attack]           else            out = in + [[out(previous) − in] * decay]           endif                        
where out(previous) represents the output from the previous sample period.
 
     The values for attack and decay are sample-rate dependent since the slew rates must be correlated to real time. The formulas for each are provided below:
 
attack=1−(1/(0.01*sampleRate))
 
decay=1−(1/(0.1*sampleRate))
 
where sample rate is in samples/second.
 
     The input to the FASD Peak Detector  3123  is always greater than or equal to zero, since it comes from the output of the absolute value function  3122 . 
     The filters  3809 - 3812  are implemented as Infinite Impulse Response (IIR) filters at a sampling frequency of 44.1 kHz. The filters are designed using the bilinear transform method. Each filter is a second order filters having one section. The filters are implemented using 32 bits fractional fixed point arithmetic. Specific formation for each filter is given in Table 1 below. In addition, the transfer functions of the filters  3810  through  3812  are shown in  FIGS. 39 through 43  respectively. The transfer function of the lowpass filter  3809  is shown in  FIG. 44 . 
     
       
         
           
               
             
               
                 TABLE 1 
               
               
                   
               
             
            
               
                 Bandpass Filters 
               
            
           
           
               
               
               
               
               
               
            
               
                 Filter 
                   
                   
                   
                   
                   
               
               
                 Frequencies 
                 −3 dB Low 
                 Center 
                 −3 dB 
                 Bandpass 
                 Bandpass 
               
               
                 (Hz) 
                 (Hz) 
                 (Hz) 
                 High (Hz) 
                 Gain 
                 Gain (dB) 
               
               
                   
               
            
           
           
               
               
               
               
               
               
            
               
                 40 
                 30 
                 38.7 
                 50 
                 1.43 
                 3.12 
               
               
                 60 
                 45 
                 58.1 
                 75 
                 1.43 
                 3.12 
               
               
                 100 
                 78 
                 96.8 
                 129 
                 1.00 
                 0.0 
               
               
                 150 
                 116 
                 145.1 
                 192 
                 1.00 
                 0.0 
               
               
                 200 
                 150 
                 193.6 
                 250 
                 0.71 
                 −2.93 
               
               
                   
               
            
           
           
               
            
               
                 Lowpass Filter 
               
            
           
           
               
               
               
               
               
            
               
                   
                 −3 dB 
                 −15 dB 
                 Bandpass 
                 Bandpass 
               
               
                   
                 (Hz) 
                 (Hz) 
                 Gain 
                 Gain (dB) 
               
               
                   
                   
               
               
                   
                 285 
                 1021 
                 1.00 
                 0.0 
               
               
                   
                   
               
            
           
         
       
     
     The Bass Control  3827  determines the amount of bass enhancement that is applied to the audio signal and provides a value between 0 and 1 to the multiplier  3826   
     The Width Control  3846  determines the amount of stereo width enhancement that is applied to the final output. The width control provides a value between to 2.82 (9 dB) to the multiplier  3845 . 
     Other Embodiments 
     The entire acoustic correction system disclosed herein may be readily implemented by software running on a DSP or personal computer, by discrete circuit components, as a hybrid circuit structure, or within a semiconductor substrate having terminals for adjustment of the appropriate external components. Adjustments by a user currently include the level of low-frequency and high-frequency energy correction, various signal-level adjustments including the level of sum and difference signals, and orientation adjustment. 
     Through the foregoing description and accompanying drawings, the present invention has been shown to have important advantages over current acoustic correction and stereo enhancement systems. While the above detailed description has shown, described, and pointed out the fundamental novel features of the invention, it will be understood that various omissions and substitutions and changes in the form and details of the device illustrated may be made by those skilled in the art, without departing from the spirit of the invention. Therefore, the invention should be limited in its scope only by the following claims.