Patent Publication Number: US-8981826-B2

Title: Semiconductor device

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a division of and claims the benefit of priority under 35 U.S.C. §120 from U.S. Ser. No. 13/915,329 filed Jun. 11, 2013, which is a division of Ser. No. 13/527,678 filed Jun. 20, 2012 (now U.S. Pat. No. 8,497,720 issued Jul. 30, 2013), which is a division of U.S. Ser. No. 13/110,971 filed May 19, 2011 (now U.S. Pat. No. 8,248,128 issued Aug. 21, 2012), which is a divisional of U.S. Ser. No. 12/986,500 filed Jan. 7, 2011 (now U.S. Pat. No. 7,973,580 issued Jul. 5, 2011), which is a divisional of U.S. Ser. No. 12/351,426 filed Jan. 9, 2009 (now U.S. Pat. No. 7,893,744 issued Feb. 22, 2011), and claims the benefit of priority under 35 U.S.C. §119 from Japanese Patent Application Nos. 2008-117491 filed Apr. 28, 2008, Japanese Patent Application No. 2008-004894 filed Jan. 11, 2008 and Japanese Patent Application No. 2008-298025 filed Nov. 21, 2008; the entire contents of each of which are incorporated herein by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     This invention relates to a semiconductor device in which a digital control power source is formed. 
     For example, PWM (Pulse Width Modulation) control in an analog power source is performed by comparison of triangle wave and feedback voltage (for example, Patent document 1 (JP-A 2001-251370 (Kokai)). In this case, a pulse width realized by analog PWM circuit can continuously change. 
     On the other hand, in a PWM circuit in a digital power source, time can be merely set discretely. When the PWM control is performed by using a clock, the clock of 1 GHz becomes required for realizing a time resolution of 1 nanosecond, and the clock of 10 GHz becomes required for realizing a time resolution of 100 picoseconds. In forming a circuit generating a clock having such a high frequency on a semiconductor substrate, a most-advanced process is required, and there is a problem that the consumption current increases for operation by the clock. 
     Moreover, conventionally, a plurality of power sources are connected in parallel. For example, when ten power sources having an output current of 10 ampere are operated in parallel, a power source having an output current of 100 ampere can be composed. 
     A current digital power IC having a parallel running function is “Master Control Architecture” composed of one master IC controlling switching of all of the phases and a plurality of driver ICs. A problem thereof is that the power system becomes failed in itself if a trouble is caused in the master IC from any cause. 
     Moreover, in Non-patent document 1 (“Current Sharing in Digitally Controlled Masterless Multi-phase DC-DC Converters”, Power Electronics Specialists Conference, 2005. PESC &#39;05. IEEE 36th), a masterless architecture in which a plurality of ICs each having capability of becoming the master are connected in parallel to realize parallel running has been disclosed. However, a master clock is input from the outside, and the current information is shared among the power ICs (the phases) through a high-speed digital bath, and therefore, the number of terminals is large and the power consumption is also large. 
     Moreover, reference voltage used for controlling the output voltage to converge to be a target value is generalized (shared) among the power source ICs. This leads to increase of the number of the terminals in actual productization, and the reference voltage-shared terminals are affected by, wiring resistance on the substrate, parasitic capacitance, and noise, and therefore, it is necessary to care for wiring among the power ICs and the design is troublesome. 
     Moreover, in the case of multi-phase operation, its interleaving setting is performed by setting outside the power ICs, and if a trouble is caused in any one of the phases from any cause, the power system becomes failed in itself, and therefore, the advantage of the masterless architecture has been lost. 
     SUMMARY OF THE INVENTION 
     According to a first aspect of the invention, there is provided a semiconductor device including: a voltage-control-type clock generation circuit having a plurality of stages of first delay elements and whose oscillation frequency is controlled according to a control voltage applied to the first delay elements; a delay circuit having a plurality of stages of second delay elements connected serially; and a selection circuit selecting one from pulse signals output by the plurality of stages of respective second delay elements, the first delay elements and the second delay elements having a same structure formed on a same semiconductor substrate, and a delay amount of the second delay elements being adjusted according to the control voltage. 
     According to a second aspect of the invention, there is provided a semiconductor device including: a voltage-control-type clock generation circuit having a plurality of stages of first delay elements and whose oscillation frequency is controlled according to a first control voltage applied to the first delay elements; a delay circuit having a plurality of stages of second delay elements connected serially; a selection circuit selecting one from pulse signals output by the plurality of stages of respective second delay elements; an error detection circuit detecting disagreement of a delay amount in the delay circuit; and an error adjustment circuit applying a second control voltage that is the first control voltage compensated based on the detection result of the error detection circuit, the first delay elements and the second delay elements having a same structure formed on a same semiconductor substrate, and a delay amount of the second delay elements being adjusted according to the second control voltage. 
     According to a third aspect of the invention, there is provided a semiconductor device including a digital PWM circuit outputting: a first pulse signal switching from a low level to a high level at a timing of being synchronized with a clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal; and a second pulse signal switching from a low level to a high level at a timing set by a shorter time resolution than that of a period of the clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal, the digital PWM circuit including: a delay circuit having a plurality of stages of delay elements connected serially; a first selection circuit selecting one from outputs of the respective delay elements and outputting a signal setting a falling timing of the first pulse signal; a second selection circuit selecting one from outputs of the respective delay elements and outputting a signal setting a rising timing of the second pulse signal; and a third selection circuit selecting one from outputs of the respective delay elements and outputting a signal setting a falling timing of the second pulse signal. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic cross-sectional view showing the circuit structure formed in a semiconductor device according to a first embodiment of the invention; 
         FIG. 2  is a circuit diagram of a first delay element shown in  FIG. 1 ; 
         FIGS. 3A to 3C  are waveform charts of main signals in a voltage-control-type clock generation circuit shown in  FIG. 1 ; 
         FIGS. 4A to 4F  are timing charts of the main signals in the circuit shown in  FIG. 1 ; 
         FIG. 5  is a schematic view showing the circuit structure formed in a semiconductor device according to a second embodiment of this invention; 
         FIG. 6  is a schematic view showing the circuit structure formed on a semiconductor device according to a third embodiment of the invention; 
         FIG. 7  is a schematic view showing the structure of an error detection circuit in  FIG. 6 ; 
         FIGS. 8A to 8C  are timing charts for each signal va, ve 1 , ve 2  shown in  FIG. 6 ,  FIG. 8A  shows the case where the delay amount varies to the direction of becoming slower,  FIG. 8B  shows the case where the delay amount varies to the direction of becoming smaller, and  FIG. 8C  shows the case where the delay amount does not vary; 
         FIG. 9  is a schematic view showing the structure of an error adjustment circuit in  FIG. 6 ; 
         FIG. 10  is a schematic view showing the circuit structure formed on a semiconductor device according to a fourth embodiment of the invention; 
         FIG. 11  is a schematic view showing the structure of the semiconductor device according to this embodiment of the invention based on a completely differential circuit as the delay element; 
         FIG. 12  is a circuit diagram of the delay element shown in  FIG. 11 ; 
         FIG. 13  is a schematic view showing a modification example without using EXOR circuit in the clock generation circuit shown in  FIG. 11 ; 
         FIG. 14  is a schematic view showing the structure of a voltage step down DC-DC converter as one example of a digital control power source according to this embodiment of the invention; 
         FIG. 15  is a circuit diagram showing one structure example of a digital PWM circuit shown in  FIG. 14  as a semiconductor device according to a fifth embodiment of the invention; 
         FIGS. 16A to 16K  are waveform charts (timing charts) of the main signals in the circuit shown in  FIG. 15 ; 
         FIG. 17  is a circuit diagram showing another structure example of the digital PWM circuit shown in  FIG. 14  as a semiconductor device according to a sixth embodiment of the invention; 
         FIG. 18  is a circuit diagram showing still another structure example of the digital PWM circuit shown in  FIG. 14  as a semiconductor device according to a seventh embodiment of the invention; 
         FIG. 19  is a circuit diagram showing still another structure example of the digital PWM circuit shown in  FIG. 14  as a semiconductor device according to an eighth embodiment of the invention; 
         FIG. 20  is a circuit diagram showing still another structure example of the digital PWM circuit shown in  FIG. 14  as a semiconductor device according to a ninth embodiment of the invention; 
         FIG. 21  is a circuit diagram showing one structure example of an output voltage control circuit in a DC-DC converter as a semiconductor device according to an twelfth embodiment of the invention; 
         FIG. 22  is a schematic view showing one example of thresholds (Bin) set in comparing the reference voltage Vref with the output voltage in the circuit in  FIG. 21 ; 
         FIG. 23  is a schematic view showing the corresponding relation between the thresholds shown in  FIG. 22  and the control parameter set in accordance with this; 
         FIG. 24  is a circuit diagram showing one structure example of an output voltage control circuit in a DC-DC converter as a semiconductor device according to an thirteenth embodiment of the invention; 
         FIG. 25  is a schematic view showing one example of thresholds (Bin) set in comparing the reference voltage Vref with the output voltage in the circuit in  FIG. 24 ; 
         FIG. 26  is a schematic view showing the corresponding relation between the thresholds shown in  FIG. 25  and the control parameter set in accordance with this; 
         FIG. 27  is a schematic view showing the structure of a digital control power source as a semiconductor device according to a fourteenth embodiment of the invention; 
         FIGS. 28A and 28B  are schematic views for describing the multiphase operation in a parallel operation power source; 
         FIG. 29  is a graph showing the relation between a current flowing through each of power ICs and the efficiency in a digital control power source as a semiconductor device according to a fifteenth embodiment of the invention; 
         FIG. 30  is a schematic view showing the structure of a digital control power source as a semiconductor device according to a sixteenth embodiment of the invention; 
         FIG. 31  is a schematic view showing the structure of a digital control power source as a semiconductor device according to a seventeenth embodiment of the invention; 
         FIG. 32  is a schematic view showing a specific example for a transmit/receive circuit of share current value information provided on each power source IC shown in  FIG. 31 ; 
         FIG. 33  is a waveform chart showing an example which each IC power source IC shown in  FIG. 31  converts its own output current value into a high-level pulse width; 
         FIG. 34  is a waveform chart for describing a wired OR function in the current share structure shown in  FIG. 31 ; 
         FIG. 35  is a waveform chart showing an example which each IC power source IC shown in  FIG. 31  converts its own output current value into a low-level pulse width; 
         FIG. 36  is a schematic view showing a structure example of each DC-DC converter shown in  FIG. 31 ; 
         FIGS. 37A to 37G  are waveform charts (timing chart) of the main signals in the circuit of  FIG. 19 ; 
         FIG. 38  is a circuit diagram showing a structure example of a digital PWM circuit in a semiconductor device according to a tenth embodiment of the invention; 
         FIGS. 39A to 39D  are waveform charts (timing chart) of the main signals in the circuit of  FIG. 38 ; and 
         FIG. 40  is a circuit diagram showing a structure example of a digital PWM circuit in a semiconductor device according to a eleventh embodiment of the invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Hereinafter, embodiments of this invention will be described with reference to drawings. 
     First Embodiment 
       FIG. 1  is a schematic cross-sectional view showing a circuit structure formed in a semiconductor device according to a first embodiment of the invention. 
     The circuit shown in  FIG. 1  is a digital PWM (Pulse Width Modulation) circuit and has a voltage-control-type clock generation circuit  10 , a delay circuit  20 , a delay element input circuit  32 , a counter  31 , a multiplexer  33 , a flipflop  34 , and so forth. These are formed on a common semiconductor substrate and composed as one semiconductor device (chip or packaged shape). 
     The voltage-control-type clock generation circuit  10  has delay elements A 0 , A 1 , A(n+1)/2, . . . An (the suffix n is odd number) serving as a plurality of stages of first delay elements. This circuit has a ring oscillator structure in which (n+1) delay elements A 0 , A 1 , A(n+1)/2, . . . An are cascade-arranged and the output of the last-stage delay element An is input to an inverter circuit  70  and the output of the inverter circuit  70  is input to the first-stage delay element A 0 . 
     An example of the circuit structure of the delay elements A 0 , A 1 , A(n+1)/2, . . . An and the inverter circuit  70  is shown in  FIG. 2 . 
     Each of the delay elements A 0 , A 1 , A(n+1)/2, . . . An has a structure in which two inverter circuits each composed of a p-type MOS transistor Mp 2  and an n-type MOS transistor Mn 1  are cascade-arranged. The gates of the transistor Mp 2  and the transistor Mn 1  are connected to each other, and the drain in which the transistors Mp 2 , Mn 1  are connected to each other serves as the output terminal, and this output terminal is connected to the gate in which the next-stage transistors Mp 2 , Mn 1  are connected to each other. 
     Furthermore, a p-type MOS transistor Mp 1  serving as a transistor controlling the operation current is connected between a power wire of a power voltage VDD and the transistor Mp 2 , and an n-type MOS transistor Mn 2  is connected between the transistor Mn 1  and a ground node. The operation current of the transistor Mp 1  is controlled by a bias voltage Vb 1 . The operation current of the transistor Mn 2  is controlled by a bias voltage Vb 2 . The inverter  80  being capable of controlling the operation current outputs a value inverted to the input. Therefore, in order to output the delayed signal without inverting the signal in each of the delay elements A 0 , A 1 , A(n+1)/2, two inverter circuits  80  are cascade-arranged. 
     The inverter circuit  70  connected to the output terminal of the last-stage delay element An has only one structure that is the same as the inverter  80  composed of the p-type MOS transistor Mp 2  and the n-type MOS transistor Mn 1 . In order that the ring oscillator oscillates, it is necessary that the stage number of the inverter circuits  80  is odd. Accordingly, by inputting the output of the last-stage delay element An to the inverter circuit  70  and inputting the output of the inverter circuit  70  to the first-stage delay element A 0 , the stage number is set to be odd. 
     By referring to  FIG. 1  again, the output of the first-stage delay element A 0  (shown in  FIG. 3(   a )) and the output of the delay element A(n+1)/2 (shown in  FIG. 3(   b )) are input to an exclusive disjunction or EXOR (EXclusive OR) circuit  11 , and the EXOR circuit  11  output a clock signal clk shown in  FIG. 3(   c ). Thereby, the clock generation circuit  10  can output a signal clk having a frequency that is two-times larger than a frequency oscillated by the plurality of delay elements A 0 , A 1 , A(n+1)/2, . . . , An and the inverter circuit  70 . 
     In the present embodiment, the clock signal clk is generated by using the EXOR circuit  11 , but another circuit except for the EXOR circuit is possible as long as outputting the frequency that is two-times larger than the frequency oscillated by the plurality of delay elements A 0 , A 1 , A(n+1)/2, . . . An, and the inverter circuit  70 . 
     The frequency clock signal clk depends on the delay amount (delay time) of each of the delay elements A 0 , A 1 , A(n+1)/2, . . . An, and the delay amount depends on the current flowing through each of the delay elements A 0 , A 1 , A(n+1)/2, . . . An. And, the current flowing each of the delay elements A 0 , A 1 , A(n+1)/2, . . . An is controlled by the control voltage Vsrc. That is, in the voltage-control-type clock generation circuit  10 , the oscillation frequency is controlled according to the applied control voltage Vsrc. 
     The delay circuit  20  also has delay elements B 0 , B 1 , B 2 , . . . Bn (the suffix n is odd) serving as a plurality of stages of second delay elements that are cascade-arranged in the same manner as the clock generation circuit  10 , and the delay elements B 0 , B 1 , B 2 , . . . Bn are connected not in a ring shape but linearly. That is, the output of the last-stage delay element Bn is not returned to the input side of the first-stage delay element B 0 . 
     The delay elements B 0 , B 1 , B 2 , . . . Bn in the delay circuit  20  have the same circuit structure as the delay elements A 0 , A 1 , A(n+1)/2, . . . An in the clock generation circuit  10  so as to be synchronized with the clock frequency (frequency of the clock signal clk). That is, the delay elements A 0 , A 1 , A(n-F 1 )/2, . . . An, and the delay elements B 0 , B 1 , B 2 , . . . Bn in the delay circuit  20  are the delay elements having the same structures formed on the same semiconductor substrate (semiconductor chip), and each of the delay elements B 0 , B 1 , B 2 , . . . Bn also have the same circuit structures as the delay elements A 0 , A 1 , A(n+1)/2, . . . An described above with reference to  FIG. 2 . 
     And, the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . Bn is controlled by the same control voltage Vsrc as applied to the clock generation circuit  10 , and therefore, the delay amount of each of the delay elements B 0 , B 1 , B 2 , . . . Bn is adjusted by the control voltage Vsrc. 
     In the former stage of the delay circuit  20 , the delay element input circuit  32  is provided, and to the delay element input circuit  32 , a count value cnt of the counter  31  and, for example, a digital signal D [MSB] with top 3 bits of a 5-bit digital signal are input. 
     Here,  FIG. 4  is a timing chart of the main signals clk, cnt, S, q 0 , q 1 , q 2 , q 3 , R, and V 0  in the circuits of  FIG. 1 . 
     The counter  31  counts up the clock signals clk oscillated by the clock generation circuit  10  one by one (by one period) as shown in  FIG. 4(   b ), and when the counted value (3 bits) cnt accords to D[MSB], the delay element input circuit  32  outputs a pulse signal va to the first-stage delay element B 0 . The pulse signal va is a pulse signal having a width of one period of the clock signal clk. 
     The example shown in  FIG. 4  indicates the case of “011”, and the pulse signal va is input to the first-stage delay element B 0  at the timing that the count value cnt becomes “011”, and the pulse signal va transmits through each of the delay elements B 0 , B 1 , B 2 , . . . Bn sequentially from the first stage, and delay according to the control voltage Vsrc is generated as shown in  FIG. 4(   d ) in the timing that each of the pulse signals q 0 , q 1 , q 2 , q 3 , . . . output by the respective delay elements B 0 , B 1 , B 2 , . . . Bn rises. 
     Each of the output q 0 , q 1 , q 2 , . . . of the respective delay elements B 0 , B 1 , B 2 , . . . Bn is input, for example, to the multiplexer  33  as a selection circuit. The multiplexer  33  selects one of the output q 0 , q 1 , q 2 , . . . of the respective delay elements B 0 , B 1 , B 2 , . . . Bn on the basis of D[LSB] with low 2 bits in the 5-bit digital signal whose top 3 bits are used as the above-described D[MSB], and the selected output is output to the reset terminal of the flipflop  34  as the reset pulse R as shown in  FIG. 4(   e ). The example shown in  FIG. 4  indicates the case that the D[LSB] has “10”, and the output q 2  is selected, and the reset pulse R synchronized with the q 2  is output to the reset terminal of the flipflop  34 . 
     To the set terminal of the flipflop  34 , the set pulse S shown in  FIG. 4(   c ) is input. The set pulse S has a predetermined period, and becomes ON, for example, when the count value cnt of the counter  31  is “000”. 
     The flipflop  34  outputs a pulse signal V 0  from the output terminal. As shown in  FIG. 4(   f ), the pulse signal V 0  switches from a low level to a high level at the rising edge of the set pulse S, and the high level is held until the reset pulse R is input, and when the reset pulse R is input, the high level switches to the low level at the rising of the reset pulse R. 
     This pulse signal V 0  becomes an output of the PWM circuit shown in  FIG. 1 , and the pulse signal V 0  is provided to the gate of the switching element (MOSFET), ON/OFF of the switching element is controlled. 
     According to this embodiment, based on the above-described set pulse S and the reset pulse R, PWM control of the pulse signal V 0  is performed. The set pulse S rises based on the count value cnt of counting up the clock signal clk one by one (period by period), and by rising of the set pulse S, the pulse signal V 0  becomes in the high level from the low level. Therefore, the rising edge of the pulse V 0  is relatively roughly determined according to what number of the clock clk the edge is in. 
     By contrast, the falling edge of the pulse signal V 0  is determined by a rising edge of any one of the outputs q 0 , q 1 , q 2 , q 3 , . . . of the respective delay elements B 0 , B 1 , B 2 , . . . Bn. Each of the rising edge of q 0 , q 1 , q 2 , q 3 , . . . is delayed with respect to the former stage by smaller time interval than one period of the clock clk, and therefore, a pulse width modulation of the pulse signal V 0  is realized by smaller time resolution than one period of the clock clk according to which (rising edge of) signal is selected out of q 0 , q 1 , q 2 , q 3 , . . . . Thereby, without requiring high-speed clock, the pulse width modulation can be realized by smaller time resolution, and increase of const or consumption current can be suppressed. 
     The delay elements A 0 , A 1 , A 2 , . . . An of the clock generation circuit  10  and the delay elements B 0 , B 1 , B 2 , B 3 , . . . Bn of the delay circuit  20  are composed by the same circuits formed on the same semiconductor substrate, and the delay amounts thereof are controlled by the same control voltage Vsrc, and therefore, resolution of accurately dividing the clock period can be simply obtained. In the circuit structure of  FIG. 1 , one clock period can be divided into about (n+1). For example, in the case of dividing it into 32, n=31 is set, and the clock signal clk is generated by the output of the delay element A 0  and the output of the delay element A 16 . As the clock period becomes longer, the transmission delay in the delay circuit  20  becomes larger, and conversely, if the clock period becomes shorter, the transmission delay in the delay circuit  20  becomes shorter. 
     Second Embodiment 
       FIG. 5  is a schematic view showing a circuit structure formed in a semiconductor device according to a second embodiment of the invention. The same signs are appended to the same components as the above-described first embodiment, and the detailed explanation thereof will be omitted. 
     In the second embodiment, the structure of the clock generation circuit is different from that of the first embodiment. The clock generation circuit  15  in the second embodiment is formed by the delay elements A 0 , A 1 , . . . A(n+1)/2 that are half of those of the clock generation circuit  10  and the inverter circuit  70 , and the clock signal clk is output from the output terminal of the inverter circuit  70 . Moreover, the EXOR circuit  11  is used in generating the clock signal clk in the first embodiment, but is not required in the second embodiment. Thereby, in this embodiment, the number of the delay elements of the clock generation circuit  15  decreases, and thereby, the occupied area in the semiconductor chip can be reduced compared to the first embodiment. 
     Third Embodiment 
       FIG. 6  is a schematic view showing a circuit structure formed on a semiconductor device according to a third embodiment of the invention. The same signs are appended to the same components as the above-described first embodiment, and the detailed explanation will be omitted. 
     Each of the clock generation circuit  10  and the delay circuit  25  have the delay elements formed on the same semiconductor substrate by the same process, and the same control voltage Vsrc is applied thereto, and therefore, in principle, by the rising edges of the outputs q 0 , q 1 , q 2 , . . . in the delay circuit  25 , one period of the clock clk is almost accurately evenly divided (delay amount per stage becomes 1/(number of the stages) of the clock period), but if the pair properties of the delay elements in both of the circuits  10 ,  25  are broken, the rising edges of the outputs q 0 , q 1 , q 2 , . . . come not to accurately evenly divide the clock period, and disagreement of the delay amount such as progress or lag of the delay occasionally caused. 
     Accordingly, in this embodiment, in addition of the above-described structure of the first embodiment, an error detection circuit  42  and an error adjustment circuit  41  are provided as shown in  FIG. 6 . Moreover, in this embodiment, a delay element B(n+1) is further additionally connected to a latter stage of the last-stage delay element Bn. The delay element B(n+1) is also formed on the same semiconductor substrate and has the same circuit structure, as the delay elements B 0 , B 1 , . . . Bn. 
     The error detection circuit  42  detects disagreement of the delay amount in the delay circuit  25  on the basis of, a signal va input to the first-stage delay element B 0 , an output signal ve 2  of the delay element B(n+1) that is a latter stage of the last stage, and an output signal ve 1  of the last-stage delay element Bn. 
     In this embodiment, in order to perform the control so that the rising edge of the output signal ve 1  of the last-stage delay element Bn is contained accurately in one clock period, the delay element B(n+1) is further added at the latter stage of the last-stage delay element Bn. 
     If the rising edge of the output signal ve 1  of the last-stage delay element Bn is not accurately contained in one clock period, the following thing is generated. 
     The case of linearly inputting a command increasing a value “1” by “1” is thought. The value D[MSB] is fixed and D[LSB] increases “1” by “1”, and the nodes are selected sequentially so as to monotonously increase like q 0 , q 1 , . . . qn. When the node of qn is output, ideally, the delay time becomes equal to one clock period. Then, in D[LSB], “+1” is added to the value of D[MSB]. In this case, in the circuit, the external one clock period+the delay time of the delay element B 0  are output. In this case, if the output of the node of qn becomes longer than one clock period, subsequently, the time output by one clock period+delay element B 0  occasionally becomes short according to the delay amount of the delay element B 0  despite increasing the value, there is possibility that the linearity is broken and that the control is largely affected. Accordingly, in this embodiment, in order to perform the control so that the rising edge of the output signal ve 1  of the last-stage delay element Bn is contained accurately in one clock period, the delay element B(n+1) is further added. 
     The structure of the error detection circuit  42  is shown in  FIG. 7 . 
     The output signal ve 1  of the last-stage delay element Bn is input to the input terminal of the flipflop  42   a , and the output signal ve 2  of the delay element B(n+1) of the latter stage of the last stage is input to the input terminal of the flipflop  42   b . And, the input signal va to the first-stage delay element B 0  is input to each of the flipflops  42   a ,  42   b  at the rising edge as a set pulse. 
     The signal va is a pulse having a width of one period of the clock clk, and as shown in  FIG. 8 , by the falling edge of the pulse va, each of the output signal ve 1  of the last-stage delay element Bn and the output signal ve 2  of the delay element B(n+1) that is the latter stage of the last stage is latched. 
     When the delay amount varies to the direction of becoming slower, both of ve 1 , ve 2  are latched to the low level by the falling edge of the signal va as shown in  FIG. 8A , and the value of “00” is output to the error adjustment circuit  41 . In this case, the error adjustment circuit  41  increases the current flowing through the delay elements B 0 , B 1 , . . . B(n+1). That is, when the current flowing through the delay elements A 0 , A 1 , A(n+1)/2, . . . An of the clock generation circuit  10  is Isrc 1  and the current Isrc 2  flowing through the delay elements B 0 , B 1 , . . . B(n+1) of the delay circuit  20  is Isrc 2 , Isrc 2  is increased more than Isrc 1  (Isrc 2 =Isrc 1 +Δi). Because the operation current increases, the delay elements B 0 , B 1 , . . . B(n+1) operates at a higher speed than the case of adding Δi. Accordingly, feedback is generated to the direction of making the delay amount smaller. 
     By contrast, when the delay amount varies to the direction of becoming smaller, both of ve 1 , ve 2  are latched to the high level by the falling edge of the signal va as shown in  FIG. 8B , and the value of “11” is output to the error adjustment circuit  41 . In this case, the error adjustment circuit  41  decreases Isrc 2  more than Isrc 1  (Isrc 2 =Isrc 1 −Δi). Because the operation current decreases, the delay elements B 0 , B 1 , . . . B(n+1) operates at a lower speed than the case of reducing Δi. Accordingly, feedback is generated to the direction of making the delay amount larger. 
     When ve 1  is latched to the high level and ve 2  is latched to the low level at the falling edge of the signal va as shown in  FIG. 8C , the delay amount is determined not to have disagreement, the value of “10” is output to the error adjustment circuit  41 . In this case, the error adjustment circuit  41  does not increase and decrease Isrc 2 , but the current Isrc 2  is held. 
     When the detection result by the error detection circuit  42  is “00” or “11”, the error adjustment circuit  41  adjusts Isrc 2  so that “00” or “11” becomes “10”. That is, the error adjustment circuit  41  applies the second control voltage Vsrc 2  that is the compensated first control voltage Vsrc 1  applied for flowing the current Isrc 1  through the delay elements A 0 , A 1 , A(n+1)/2, . . . An, to the delay circuit  25 , and thereby, the current Isrc 2  flows through the delay elements B 0 , B 1 , . . . B(n+1) of the delay circuit  25  by the second control voltage Vsrc 2   
     One example of the circuit structure of the error adjustment circuit  41  is shown in  FIG. 9 . 
     The error adjustment circuit  41  has a decoder  43 , for example eight switching elements (N-type MOS) M 1  to M 8  ON/OFF-controlled with receiving the signal from the decoder  43  to the gates thereof, and a current mirror circuit composed of P-type MOSs  51 ,  53  and N-type MOSs  52 ,  54 . 
     In the initial state, all of the output ports i 1  to i 8  of the decoder  43  output “0”, and all of the eight switching elements M 1  to M 8  are set to be in the OFF state, and accordingly, the current does not flow through the lines L 1  and L 2 . In this case, P-type MOS  51  is set to be in the ON state by the first control voltage Vsrc 1  applied to the gate thereof, and thereby the same current flows through the nodes n 1 , n 2 . The bias voltage corresponding to this current is applied to the gate of the P-type MOS  55  corresponding to each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1), and the P-type MOS  55  becomes in the ON state, and the current is supplied to each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1). 
     When the detection result in the above-described error detection circuit  42  is “00”, first, “1” is output only from the port i 5  (the other ports i 1  to i 4  and i 6  to i 8  remain “0”.). When the port i 5  becomes “1”, the switching element M 5  becomes ON, the current flows through the line L 2 . When the current flowing through the line L 2  is I 1  and the current flowing through the node n 2  in the above-described initial state is I, the current of I+I 1  flows through the node n 2 . And, the gate of the P-type MOS  55  is biased according to the current of I+I 1 , and as a result, a larger current than that of the initial state flows through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1), and the delay amount is compensated to the direction of becoming smaller. 
     After the compensation, if the detection result in the error detection circuit  42  still remains “00”, “1” is also output from the port i 6  in addition of the port i 5 . When the ports i 5  and i 6  become “1”, the switching elements M 5  and M 6  become ON, the current two-times larger than I 1  flows through the line L 2 . Accordingly, the current of I+(2×I 1 ) flows through the node n 2 , and the gate of the P-type MOS  55  is biased according to the current of I+(2×I 1 ), and the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1) is adjusted to be larger, and the delay amount is compensated to the direction of becoming smaller. 
     When the detection result in the error detection circuit  42  becomes “10”, the above-described current adjustment in the error adjustment circuit  41  is finished, but if the detection result in the error detection circuit  42  still indicates “00”, by sequentially setting the ports i 7 , i 8  to be “1”, the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1) is adjusted so as to be larger. 
     When the detection result in the above-described error detection circuit  42  is “11”, first, “1” is output only from the port i 1  (the other ports i 2  to i 4  and i 5  to i 8  remain “0”.). When the port i 1  becomes “1”, the switching element M 1  becomes ON, the current flows through the line L 1 . When the current flowing through the line L 1  is I 1  and the current flowing through the node n 1  in the above-described initial state is I, the current of I−I 1  flows through the node n 1 . And, the gate of the P-type MOS  55  is biased according to the current of I−I 1 , and as a result, a smaller current than that of the initial state flows through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1), and the delay amount is compensated to the direction of becoming slower. 
     After the compensation, if the detection result in the error detection circuit  42  still remains “11”, “1” is also output from the port i 2  in addition of the port i 1 . When the ports i 1  and i 2  become “1”, the switching elements M 1  and M 2  become ON, the current two-times larger than I 1  flows through the line L 1 . Accordingly, the current of I−(2×I 1 ) flows through the node n 1 , and the gate of the P-type MOS  55  is biased according to the current of I−(2×I 1 ), and the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1) is adjusted to be smaller, and the delay amount is compensated to the direction of becoming slower. 
     When the detection result in the error detection circuit  42  becomes “10”, the above-described current adjustment in the error adjustment circuit  41  is finished, but if the detection result in the error detection circuit  42  still indicates “11”, by sequentially setting the ports i 3 , i 4  to be “1”, the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1) is adjusted so as to be smaller. 
     As described above, by adjusting the current flowing through each of the delay elements B 0 , B 1 , B 2 , . . . B(n+1) on the basis of the detected disagreement of the delay amount to compensate the disagreement of the delay amount, it can be realized that one period of the clock clk is almost accurately evenly divided by rising edges of the outputs q 0 , q 1 , q 2 , . . . (delay amount per stage becomes 1/(number of the stages) of the clock period). As a result, the precise and reliability of the PWM control can be improved. 
     In the above-described specific example, four-stage adjustment can be performed for both of the directions of making the delay smaller and slower, but here, the assumed disagreement of the delay amount is only based on variation of the semiconductor process, and considering the current semiconductor process, the disagreement of the delay amount cannot be thought to be too large, and the disagreement can be sufficiently addressed by the adjustment widths of several stages (several bits) as shown in  FIG. 9 , and thereby, cost up beyond necessity can be suppressed. 
     Fourth Embodiment 
       FIG. 10  is a schematic view showing a circuit structure formed on a semiconductor device according to a fourth embodiment of the invention. The same signs are appended to the same components as the above-described embodiments, and the detailed explanation will be omitted. 
     In the fourth embodiment, the structure of the clock generation circuit is different from that of the third embodiment. The clock generation circuit  15  in the fourth embodiment is formed by the delay elements A 0 , A 1 , . . . A(n+1)/2 that are half of those of the clock generation circuit  10  in the third embodiment and the inverter circuit  70 , and the clock signal clk is output from the output terminal of the inverter circuit  70 . Moreover, the EXOR circuit  11  is used in generating the clock signal clk in the third embodiment, but is not required in the fourth embodiment. Thereby, in this embodiment, the number of the delay elements of the clock generation circuit  15  decreases, and thereby, the occupied area in the semiconductor chip can be reduced compared to the third embodiment. 
     In the above-described embodiments, one-input-and-one-output circuit is used as the delay element, but, a completely differential circuit may be used as shown in  FIGS. 11 ,  12 . 
     In  FIG. 11 , the clock generation circuit  16  has (n+1) delay elements C 0 , C 1 , . . . Cn. In the above-described embodiments, the inverter circuit  70  is connected to the latter stage of the last-stage delay element, and by contrast, in the structure shown in  FIG. 11 , without using the inverter  70 , the ring oscillator structure is formed by (n+1) delay elements C 0 , C 1 , . . . Cn. 
     In each of the first-stage C 0  to (n−1) stage in the delay elements, (+) output terminal is connected to (−) input terminal of the next stage, and (−) output terminal is connected to (+) input terminal of the next stage, but for generating oscillation, the (+) output terminal of the last-stage Cn is connected to (+) input terminal of the first-stage C 0 , and the (−) output terminal thereof is connected to (−) input terminal of the first-stage C 0 . 
     The delay circuit  26  also has (n+1) delay elements C 0 , C 1 , . . . Cn cascade-arranged in the same manner as the clock generation circuit  16 , but the delay elements C 0 , C 1 , . . . Cn are connected not in a ring shape but linearly. 
     The delay elements C 0 , C 1 , . . . Cn in the delay circuit  26  have the same circuit structures as the delay elements C 0 , C . . . , . . . Cn of the clock generation circuit  16  so as to being synchronized with the clock period in the clock generation circuit  16 . That is, the delay elements of the clock generation circuit  16  and the delay elements of the delay circuits  26  are the same structures formed on the same semiconductor substrate (semiconductor chip). And, the current flowing through each of the delay elements C 0 , C 1 , . . . Cn in the delay circuit  26  is controlled by the same control voltage Vsrc as applied to the clock generation circuit  16 , and the delay amount is adjusted. 
     The flipflop  34  operates by detecting the rising edge of a pulse, and therefore, it is necessary that the delay circuit  26  also alternately outputs “+” and “−” as the two outputs of each of the delay elements C 0 , C 1 , . . . Cn so that the rising edge is output. 
     One example of the circuit structure of each of C 0 , C 1 , . . . Cn composed as the completely differential circuit is shown in  FIG. 12 . 
     The p-type transistors Mp 1 , Mp 2  have the common gate and the common source potential (power voltage VDD) and connected respectively to drains of n-type MOS transistors Mn 1 , Mn 2 . Moreover, the sources of the n-type MOS transistors Mn 1 , Mn 2  are connected to the drain of n-type MOS transistor Mn 3 . 
     The gate of the transistor Mn 1  functions as (+) input terminal, and the gate of the transistor Mn 2  functions as (−) input terminal. The drains of the transistor Mp 1  and the transistor Mn 1  are connected to each other to be (−) output terminal. The drains of the transistor Mp 2  and the transistor Mn 2  are connected to each other to be (+) output terminal. 
     By controlling the gate potential Vb 1  of the transistors Mp 1 , Mp 2  and the gate potential Vp 2  of the transistor Mn 3 , the operation current of this current can be adjusted, and the delay time can be controlled. 
     With referring to  FIG. 11  again, the (+) terminal output of the first-stage delay element C 0  and the (+) terminal output of the delay element C(n+1)/2 are input to the EXOR circuit  11 , and the EXOR circuit  11  outputs the clock signal clk. 
     Because one clock period is determined by the delay time of (n+1) delay elements and the delay time of the inverter circuit  70 , one clock period cannot be accurately divided into 1/(n+1) by the influence of the delay time of the inverter circuit  70 , and by contrast, in the structure of  FIG. 11 , the oscillation is performed by (n+1) delay elements C 0 , C 1 , Cn, and therefore, one period can be accurately divided to 1/(n+1). 
     Moreover, in the structure of  FIG. 11 , it is also possible that without using EXOR circuit  11 , the clock generation circuit is composed only by C(n+1)/2 delay elements C 0 , C 1 , . . . C(n+1)/2 as shown in  FIG. 13 . In the clock generation circuit  17 , by decreasing the number of the delay elements to be half in comparison to the clock generation circuit shown in  FIG. 11 , the occupied area in the semiconductor chip can be reduced. 
     The application example of the digital PWM circuit of each of the above-described embodiments includes a digital control power source. Hereinafter, a DC-DC converter will be exemplified as the digital control power source. 
       FIG. 14  is a schematic view showing a structure example of the DC-DC converter, and the DC-DC converter has switching elements Q 1 , Q 2 , an inductor L, a condenser C, a control circuit switching ON/OFF of the switching elements Q 1 , Q 2 , and so forth. 
     The DC-DC converter is a voltage step down DC-DC converter (buck converter) for obtaining (average) output voltage Vout lower than the input voltage Vin by alternately switching ON/OFF of the high-side switching element Q 1  and the low-side switching element Q 2 . 
     The elements (a comparator  3 , an A/D conversion circuit  4 , a PID (proportional-integral-derivative) compensator  5 , a digital PWM circuit  6 , and the switching elements Q 1 , Q 2 ) surrounded by dashed line in  FIG. 14  are composed as IC 30  that is one chip (or one package). 
     Each of the switching elements Q 1 , Q 2  is, for example, power MOSFET (Metal-Oxide-Semiconductor Field Effect Transistor), and each of the gate terminals is connected to the digital PWM circuit  6 . 
     The source-drain of the high-side switching element Q 1  is connected between the input voltage source  1  and the switch node  40 , and the source-drain of the low-side switching element Q 2  is connected between the switch node  40  and the ground. The switch node  40  that is a connective point of both of the switching elements Q 1 , Q 2  is connected to a load  2  through the inductor L. Moreover, as a filter element for not largely changing the output voltage in a short time, a condenser C is connected between the output side of the inductor L and the ground. 
     For controlling ON/OFF of the switching elements Q 1 , Q 2 , the switching pulse having an almost inverted phase is provided from the digital PWM circuit  6  to the gate of each of the switching elements Q 1 , Q 2 . 
     When the high-side switching element Q 1  is ON and the low-side switching element Q 2  is OFF, the current flows from the input voltage source  1  through the switching element Q 1  to the inductor L, and the inductor current increases, and energy is accumulated in the inductor L. And, when the high-side switching element Q 1  is OFF and the low-side switching element Q 2  is ON, the inductor L releases the accumulated energy, and a return current flowing from the ground through the switching element Q 2  to inductor L flows (the inductor current comes to decrease). 
     The output voltage Vout is controlled so as to converge to a target output voltage (reference voltage Vref). Specifically, the output voltage Vout is input to the comparator  3 , and the comparator  3  outputs the comparison result of the output voltage Vout and the reference voltage Vref to the A/D conversion circuit  4 , and the A/D conversion circuit  4  outputs an error signal e indicating how much the actual output voltage disagrees with the target output voltage, as a digital signal to the compensator  5 . By receiving this error signal e, the compensator  5  calculates duty of ON/OFF of the switching elements Q 1 , Q 2  and output the duty as a duty command value d to the digital PWM circuit  6 . 
     Moreover, when both of the switching elements Q 1 , Q 2  are simultaneously set to be in the ON state, very large current (pass-through current) comes to flow through both of the switching elements Q 1 , Q 2  to the ground. For avoiding this, in setting the duty of ON/OFF of the switching elements Q 1 , Q 2 , the dead time that is a period in which both of the switching elements Q 1 , Q 2  are OFF is set. 
     Hereinafter, with reference to  FIGS. 15 to 20  and  38  to  40 , specific examples of the digital PWM circuit will be explained. Each of the digital PWM circuits  81  to  87  shown as follows corresponds to the digital PWM circuit  6  shown in  FIG. 14 . 
     Fifth Embodiment 
       FIG. 15  is a schematic view illustrating a structure of a digital PWM circuit  81  according to a fifth embodiment of the invention. 
     The digital PWM circuit  81  generates gate control signals according to the duty command value d transmitted from the compensator  5 . The gate control signals Out_mx 2  and dl 1 _clk operate flipflop  71  and generate gate signals for switching ON/OFF of the high-side switching element Q 1 . The gate control signals Out_mx 3  and Out_mx 4  operate the flipflop  72  and generate gate signals for switching ON/OFF of the switching element Q 2 . 
     Moreover, the digital PWM circuit  81  further has a clock generation circuit, which is not shown. This clock generation circuit has a ring oscillator structure in which a plurality of delay elements are cascaded-arranged in a ring shape in the same manner as above-described embodiments, and the digital PWM circuit  81  operates with being synchronized with the clock signal (clock) generated by the clock generation circuit. Moreover, the delay elements in the clock generation circuit and the delay elements D of each of the delay circuits  61  to  63  shown in  FIG. 15  are formed to have the same structures on the same semiconductor substrate. 
       FIG. 16  is a timing chart of main signals in the digital PWM circuit  81 . 
     The counter  46  counts clock ( FIG. 16  ( a )) generated by the clock generation circuit one by one (period by period) as shown in  FIG. 16(   b ), and supplies the clock to each of the comparators  48 ,  49 ,  57 ,  58 , for example, as the 3-bit count values. 
     Upper several bits (MSB) out of the duty command value (digital signal) d calculated in the compensator  5  shown in  FIG. 14  are supplied to the comparators  48 ,  57 ,  58  through the input resistor  47 . Lower several bits (LSB) out of the duty command value d are supplied through the input resistor  47  to multiplexers MUXA, MUXB, and MUXC that are selection circuits for plural inputs and one output. 
     The comparator  48  outputs the signal dl 2 _clk ( FIG. 16(   d )) having a pulse width of one clock period to the delay circuit  61  at the timing that the count value agrees with MSB. This signal dl 2 _clk transmits through each of the delay elements D sequentially from the first stage, and delay is generated in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D is input to the multiplexer MUXA. The multiplexer MUXA selects one ( FIG. 16(   e )) of output signals of the respective delay elements on the basis of the signal LSB and outputs the one to the reset terminal R of the flipflop  71 . 
     The comparator  49  outputs the signal dl 1 _clk ( FIG. 16(   c )) having a pulse width of one clock period to the set terminal S of the flipflop  71  at the timing that the count value agree with, for example, “000”. 
     The flipflop  71  outputs the pulse signal shown in  FIG. 16(   f ) from the output terminal Q. This output signal switches from the low level to the high level at the timing that the signal dl 1 _clk ( FIG. 16(   c ) input to the set terminal S switches from the low level to the high level, and the high level is held while the output signal Out_mx 2  ( FIG. 16(   e )) of the multiplexer MUXA input to the reset terminal R is in the low level, and the high level switches to the low level at the timing that the signal Out_mx 2  input to the reset terminal R switches from the low level to the high level. The output signal ( FIG. 16(   f )) of the flipflop  71  is supplied to control terminal (gate) of the high-side switching element Q 1 , and based thereon, ON/OFF of the switching element Q 1  is switched. 
     That is, the rising timing of the switching pulse for switching ON/OFF of the high-side switching element Q 1  is set at the timing of being synchronized with clock signal (clock), and the falling timing is set by the smaller time resolution than that of the clock signal (clock) by utilizing a signal delay in the delay circuit  61 . 
     For the switching pulse for switching ON/OFF of the low-side switching element Q 2 , both of rising and falling timings are set by smaller time resolution than that of the clock signal (clock) by using the delay circuits  62 ,  63  in order to enhance precision of the above-described dead time adjustment. 
     That is, the comparator  57  outputs the signal dl 3 _clk ( FIG. 16(   i )) having a pulse width of one clock period to the delay circuit  62  at the timing that the count value agrees with MSB. This signal dl 3 _clk transmits through each of the delay elements D sequentially from the first stage, and delay is generated in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D in the delay circuit  62  is input to the multiplexer MUXB. The multiplexer MUXB selects one ( FIG. 16(   j )) of output signals of the respective delay elements on the basis of the signal LSB and outputs the one to the reset terminal R of the flipflop  72 . 
     The comparator  58  outputs the signal dl 4 _clk ( FIG. 16(   g )) having a pulse width of one clock period to the delay circuit  63  at the timing that the count value agrees with MSB. This signal dl 4 _clk transmits through each of the delay elements D sequentially from the first stage, and delay is generated in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D in the delay circuit  63  is input to the multiplexer MUXC. The multiplexer MUXC selects one ( FIG. 16(   h )) of output signals of the respective delay elements D on the basis of the signal LSB and outputs the one to the set terminal S of the flipflop  72 . 
     The flipflop  72  outputs the pulse signal shown in  FIG. 16(   k ) from the output terminal Q. This output signal switches from the low level to the high level at the timing that the output signal out_mx 4  ( FIG. 16(   h ) of the multiplexer MUXC input to the set terminal S switches from the low level to the high level, and the high level is held while the output signal Out_mx 3  ( FIG. 16(   j )) of the multiplexer MUXB input to the reset terminal R is in the low level, and the high level switches to the low level at the timing that the signal Out_mx 3  input to the reset terminal R switches from the low level to the high level. The output signal ( FIG. 16(   k )) of the flipflop  72  is supplied to control terminal (gate) of the low-side switching element Q 2 , and based thereon, ON/OFF of the switching element Q 2  is switched. 
     Sixth Embodiment 
       FIG. 17  is a schematic view illustrating a structure of a digital PWM circuit  82  according to a sixth embodiment of the invention. The same signs are appended to the same components as the components shown in  FIG. 15 . 
     In this embodiment, the functions of the clock generation circuit and three delay circuits  61  to  63  in the above-described circuit shown in  FIG. 15  is consolidated to one. That is, the delay circuit  64  also has a function of generating the clock signal (clock). 
     The delay circuit  64  has a plurality of cascade-arranged delay elements D and an inverter circuit  65 , and has a ring oscillator structure in which the output of the last-stage delay element D is input to the inverter circuit  65  and the output of the inverter  65  is input to the first-stage delay element D. 
     Moreover, in this embodiment, an AND gate  66  having the output of comparator  48  and the output of the multiplexer MUXA as two inputs, an AND gate  67  having the output of comparator  58  and the output of the multiplexer MUXB as two inputs, and an AND gate  68  having the output of comparator  57  and the output of the multiplexer MUXC as two inputs, are provided. The output of the AND gate  66  is input to the reset terminal R of the flipflop  71 , and the output of the AND gate  67  is input to the reset terminal R of the flipflop  72 , and the output of the AND gate  68  is input to the set terminal S of the flipflop  72 . 
     The functions as the entire digital PWM circuit are the same as the circuit shown in  FIG. 15 , but in this embodiment, when the count value input to the comparator  48  agrees with MSB, the output of the corresponding multiplexer MUXA is set to be effective, and when the count value input to the comparator  58  agrees with MSB, the output of the corresponding multiplexer MUXB is set to be effective, and when the count value input to the comparator  57  agrees with MSB, the output of the corresponding multiplexer MUXC is set to be effective. 
     That is, at the timing that both of two inputs of AND gate  66  become in the high level, the reset signal is transmitted to the reset terminal R of the flipflop  71 , and the output of the flipflop  71  switches from the high level to the low level. Moreover, at the timing that both of two inputs of AND gate  68  become in the high level, the set signal is transmitted to the set terminal S of the flipflop  72 , and the output of the flipflop  72  rises from the low level to the high level, and at the timing that both of two inputs of AND gate  67  become in the high level, the reset signal is transmitted to the reset terminal R of the flipflop  72 , and the output of the flipflop  72  switches from the high level to the low level. 
     As described above, also in this embodiment, the rising timing of the switching pulse for switching ON/OFF of the high-side switching element Q 1  is set at the timing of being synchronized with the clock signal (clock), and the falling timing thereof is set by the smaller time resolution than that of the clock signal (clock), and for the switching pulse for switching ON/OFF of the low-side switching element Q 2 , both of the rising and falling timings are set by the smaller time resolution than that of the clock signal (clock). 
     For the above-described delay circuits (delay lines)  61 ,  62 ,  63  in the embodiment shown in  FIG. 15 , it is required that each of the entire delay amounts thereof is equal to one period of the clock signal (clock) and that the delay amounts of the respective delay elements D are equal. Currently, it is difficult to satisfy this, and in the structure of  FIG. 15 , the area occupied in one chip tends to be large. 
     By contrast, in the embodiment shown in  FIG. 17 , the functions corresponding to the clock generator and the delay circuits  61  to  63  are consolidated to one circuit  64 , and thereby, the entire delay amount can be precisely set to be one clock period, and the semiconductor device can be composed by the small number of elements, and therefore, variation of the delay amounts of the respective delay elements is also suppressed, and the occupied area of the delay elements is also reduced. That is, in this embodiment, by simplifying the structure, the improvement of the characteristics and reduction of the area can be achieved. 
     Seventh Embodiment 
       FIG. 18  is a schematic view illustrating a structure of a digital PWM circuit  83  according to a seventh embodiment of the invention. The same signs are appended to the same components as the components shown in  FIG. 17 . 
     In this embodiment, the delay circuit  64  is incorporated as one component of DLL (Delay Locked Loop) or PLL (Phase Locked Loop). In this DLL/PLL  75 , the PD (Phase Detector)  74  detects the phase difference between the external clock and the internal clock (clock) generated by the delay circuit  64  and feedbacks the phase difference to the delay circuit  64  and synchronizes the internal clock (clock) with the external clock, and thereby, the phase (frequency) of the internal clock (clock) can be maintained to be constant without depending on change of treatment, voltage, temperature, or the like. The system of the DC-DC converter is operated with being synchronized with the internal clock (clock), and thereby, characteristic improvement of the entire system can be achieved. 
     Eighth Embodiment 
       FIG. 19  is a schematic view illustrating a structure of a digital PWM circuit  84  according to an eighth embodiment of the invention. The same signs are appended to the same components as the above-described embodiment. Moreover,  FIG. 37  is a timing chart of main signals in the digital PWM circuit  84 . 
     In the same manner as the above-described embodiments, to the reset terminal R of the flipflop  71  outputting the high-side switching pulse, an output signal Out_mx 2  ( FIG. 37(   b )) of the multiplexer MUXA is input, and to the set terminal S thereof, the output signal dl 1 _clk ( FIG. 37(   a )) of the comparator  49  is input. 
     To the set terminal S of the flipflop  72  outputting a low-side switching pulse, the signal ( FIG. 37(   e )) delaying the signal Out_mx 2  by the delay element  78  is input, and to the reset terminal R thereof, the signal ( FIG. 37(   f )) delaying the signal dl 1 _clk by the delay element  77  is input. The delay amounts of the delay element  77  and the delay element  78  can be adjusted. 
     Moreover, the output signal ( FIG. 37(   c )) of the high-side flipflop  71  is delayed by the delay element  76  ( FIG. 37(   d )). 
     In this embodiment, only one delay circuit  61  determining the time resolution of the digital PWM is used, and the dead time is produced by utilizing signal delay by the delay elements  76  to  78  provided separately from the delay circuit  61 . Specifically, by the difference between the delay amount of the delay element  76  and the delay amount of the delay element  77 , the low-side switching element Q 2  becomes OFF, and the dead time is generated until the high-side switching element Q 1  becomes ON, and by the difference between the delay amount of the delay element  76  and the delay amount of the delay element  78 , the high-side switching element Q 1  becomes OFF, and the dead time is generated until the low-side switching element Q 2  becomes ON. The delay elements  76  to  78  are not required to have the same characteristics as the delay circuit  61 , and therefore, the design thereof is easy. 
     Ninth Embodiment 
     Like a digital PWM circuit  85  shown in  FIG. 20 , by using DLL/PLL  75  for the embodiment shown in  FIG. 19 , the internal clock (clock) can be synchronized with the external clock and maintained to be constant. When the system of the DC-DC converter is operated with being synchronized with the internal clock (clock), the characteristic improvement of the entire system can be achieved. 
     Tenth Embodiment 
       FIG. 38  is a circuit diagram illustrating a structure of a digital PWM circuit  86  according to a tenth embodiment of the invention. This digital PWM circuit  86  has the same structure as the digital PWM circuit  81  in the fifth embodiment shown in  FIG. 15 . 
     Moreover,  FIG. 39  is a timing chart of main signals in the digital PWM circuit  86  of this embodiment. 
     D 1  of  FIG. 39(   a ) represents a gate driving signal of the high-side switching element Q 1  in the same manner as above-described  FIG. 16(   f ), and D 2  of  FIG. 39(   d ) represents a gate driving signal of the low-side switching element Q 2  in the same manner as  FIG. 16(   k ). 
     The D 2 _set of  FIG. 39(   b ) represents the output of the multiplexer  102  in the  FIG. 38 , and the D 2 _set corresponds to the out_mx 4  ( FIG. 16(   h )) of the multiplexer MUXC in the circuit of  FIG. 15 . D 2 _reset of  FIG. 39(   c ) represents the output of the multiplexer  103  in the circuit of  FIG. 38 , and the D 2 _reset corresponds to the output Out_mx 3  ( FIG. 16(   j )) of the multiplexer MUXB in the circuit of  FIG. 15 . 
     Moreover, in  FIG. 39 , “duty” represents ON time in one period (one switching cycle) of D 1 , and “td 1 ” represents a dead time after D 1  becomes OFF from ON until D 2  becomes ON from OFF, and “td 2 ” represents a dead time after D 2  becomes OFF from ON until D 1  becomes ON from OFF. 
     The digital PWM circuit  86  according to the tenth embodiment shown in  FIG. 38  has three delay circuits  91  to  93  and three multiplexers  101  to  103  provided so as to correspond to the delay circuits respectively. 
     Each of the delay circuits  91  to  93  has a plurality of stages of cascade-arranged delay elements D. Each of the delay elements D has the same circuit structure formed on the same semiconductor substrate. Moreover, in the delay circuit  92  and the delay circuit  93 , the numbers (stage number) of the delay elements D are the same, and the control current Isrc 2  supplied to each of the delay elements D is also the same. 
     In the former stages of the respective delay circuits  91  to  93 , comparators  48 ,  57 ,  58  are provided, and to each of the comparators  48 ,  57 ,  58 , the count value cnt of the counter  46  and, for example, a digital signal D [MSB] with top 3 bits of a 5-bit digital signal are input. 
     The counter  46  counts the clock signal clk ( FIG. 16(   a )) one by one (period by period) as shown in  FIG. 16(   b ), and supplies the signal to each of the comparators  48 ,  57 ,  58  as, for example, 3-bit count value. 
     At the timing that the count value agrees with the MSB[duty], the comparator  48  outputs the signal va (corresponding to dl 2 _clk of  FIG. 16(   d )) having a pulse width of one clock period, to the delay circuit  91 . The signal va transmits through each of the delay elements D sequentially from the first stage and generates delay in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D in the delay circuit  91  is input to multiplexer  101 . The multiplexer  101  selects one from the output signals of the respective delay elements D on the basis of the digital signal LSB[duty] and output the one to the reset terminal R of the flipflop  71 . The reset pulse corresponds to Out_mx 2  of  FIG. 16(   e ). 
     Moreover, to the set terminal of the flipflop  71 , the set pulse (corresponding to dl 1 _clk of  FIG. 16(   c )) is input at the timing of being synchronized with the clock signal. 
     The flipflop  71  outputs the signal D 1  (corresponding to signal of  FIG. 16(   f )) shown in  FIG. 39(   a ) from the output terminal Q. This output signal switches from the low level to the high level at the timing that the set pulse switches from the low level to the high level, and the high level is held while the reset pulse (output of the multiplexer  101 ) is in the low level, and the high level switches to the low level at the timing that the reset pulse switches from the low level to the high level. The output signal D 1  of the flipflop  71  is supplied to control terminal (gate) of the high-side switching element Q 1 , and based thereon, ON/OFF of the switching element Q 1  is switched. 
     That is, the rising timing of the switching pulse for switching ON/OFF of the high-side switching element Q 1  is set at the timing of being synchronized with clock signal (clock), and the falling timing is set by the smaller time resolution than that of the clock signal by utilizing a signal delay in the delay circuit  91 . 
     For the switching pulse for switching ON/OFF of the low-side switching element Q 2 , both of rising and falling timings are set by smaller time resolution than that of the clock signal by using the delay circuits  92 ,  93  in order to enhance precision of the above-described dead time adjustment. 
     That is, the comparator  57  outputs the signal vb having a pulse width of one clock period to the delay circuit  92  at the timing that the count value agrees with MSB [duty+td 1 ]. This signal vb transmits through each of the delay elements D sequentially from the first stage, and delay is generated in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D in the delay circuit  92  is input to the multiplexer  102 . The multiplexer  102  selects one of output signals of the respective delay elements D on the basis of the signal LSB [duty+td 1 ] and outputs the one to the set terminal S of the flipflop  72  as the set pulse D 2 _set ( FIG. 39(   b )). 
     The comparator  58  outputs the signal vc having a pulse width of one clock period to the delay circuit  93  at the timing that the count value agrees with MSB [1−td 2 ]. This signal vc transmits through each of the delay elements D of the delay circuit  93  sequentially from the first stage, and delay is generated in the rising timing of the output signal of each of the delay elements D. 
     The output signal of each of the delay elements D in the delay circuit  93  is input to the multiplexer  103 . The multiplexer  103  selects one of output signals of the respective delay elements on the basis of the signal LSB [1−td 2 ] and outputs the one to the reset terminal R of the flipflop  72  as the reset pulse D 2 _reset ( FIG. 39(   c )). 
     The flipflop  72  outputs the signal D 2  shown in  FIG. 39(   d ) from the output terminal Q. This output signal D 2  switches from the low level to the high level at the timing that the set pulse D 2 _set switches from the low level to the high level, and the high level is held while the reset pulse D 2 _reset is in the low level, and the high level switches to the low level at the timing that the reset pulse D 2 _reset switches from the low level to the high level. The output signal D 2  of the flipflop  72  is supplied to control terminal (gate) of the low-side switching element Q 2 , and based thereon, ON/OFF of the switching element Q 2  is switched. 
     For enhancing voltage precision of the output voltage in the DC-DC converter, the driving signal D 1  of the high-side switching element Q 1  is required for being generated in a high time resolution. Therefore, for the delay elements D composing the delay circuit  91  setting the falling timing of D 1 , it is required to supply the control current Isrc 1  that is large to some extent in order to shorten the delay time thereof. 
     Moreover, for shortening the time resolution with respect to the clock period, the number of the delay elements D becomes larger, and when there are three such delay circuits, the occupied area of the digital PWM circuit in the chip becomes large. 
     On the other hand, in generating the driving signal D 2  of the low-side switching element Q 2 , the time resolution being as high as that of the high-side driving signal D 1  is not required. For example, if the time resolution of the high-side driving D 1  is 100 picoseconds, the time resolution of the low-side driving signal D 2  is sufficient to be about 1 nanosecond, which is ten-times lower. This is because even if the ON/OFF timing of the low-side switching element Q 2  is changed by a unit of 100 picoseconds, conversion efficiency of the power is not affected. 
     Accordingly, in this embodiment, when the time resolution of the driving signal D 1  is dt 1  as shown in  FIG. 39 , the time resolution dt 2  of each of the D 2 _set signal determining the rising timing of the driving signal D 2  and the D 2 _reset signal determining the falling timing thereof is set to be longer than dt 1  (dt 1 &lt;dt 2 ). 
     Specifically, in the circuit of  FIG. 38 , the current Isrc 2  supplied to each of delay elements D in delay circuits  92 ,  93  is decreased to be smaller than the current Isrc 1  supplied to each of delay elements D of the delay circuit  91  (Isrc 1 &gt;Isrc 2 ). For example, when n is an integer of 1 or more, the relation of Isrc 1 =Isrc 2 ×2 n  can be provided. However, in this case, the relation that the delay time of each of delay elements D is proportional to the current supplied to each of the delay elements D is established. As the current supplied to the delay elements is smaller, the time for charging the gate capacity is required more by the amount thereof, and the delay time per delay element is longer, and the dt 1 &lt;dt 2  can be realized. Moreover, to reduce the current supplied to the delay circuits  92 ,  93  is to be capable of making the consumption current smaller. 
     As described above, the delay time becomes longer as the current flowed through the delay elements is set to be smaller, but when the delay time becomes long, there is danger that the total delay amount by the entire delay elements comes not to be contained in one clock period, and therefore, in this embodiment, the numbers (stage numbers) of the delay elements in the delay circuits  92 ,  93  is decreased to be smaller than the number (stage number) of the delay elements of the delay circuit  91 . That is, when the number of the delay elements of the delay circuit  91  is (k+1) and the number of delay elements in each of the delay circuits  92 ,  93  is (m+1), k&gt;M is established. Furthermore, when n is an integer of 1 or more, the relation of (k+1)=(m+1)×2 n  can be provided. By reducing the number of the delay elements, not only the occupied area of the delay elements can be reduced but also the circuit scale of the multiplexers  102 ,  103  receiving the outputs of the respective delay elements can be smaller. 
     That is, according to this embodiment, without affecting the power conversion efficiency very much, cost down by reduction of power consumption or reducing of circuit scale can be achieved. 
     In the above explanation, in the delay circuit  92  and the delay circuit  93 , the number of the delay elements is set to be the same m, and the supplied current is set to be the same Isrc 2 , but by the two delay circuits  92 ,  93 , the number of the elements or the supplied current may be set to be different. 
     In  FIG. 36 , in the regenerative period in which the load current iL is small and the high-side switching element Q 1  is OFF and the low-side switching element Q 2  is ON, there is a mode that the magnetic energy of the inductor L becomes 0 at a certain time and that the current flows from the output terminal through the low-side switching element Q 2  to the ground. In this case, not only the charge is lost from the condenser in the output terminal but also conduction loss is caused by ON resistance of Q 2  by the current flowing through the switching element Q 2 . Thereby, the conversion efficiency of the power is significantly degraded. 
     Accordingly, it is desirable that in the period that the current flows from the output terminal through the low-side switching element Q 2  to the ground, Q 2  is set to be OFF. By setting Q 2  to be OFF, the loss is not caused because the path of the current does not exist, the conversion efficiency of the power can be improved. 
     It can be easily thought that for the timing of setting Q 2  to be OFF, sufficient time before the magnetic energy of the inductor L becomes 0 is taken for setting OFF. In this case, because the magnetic energy of the inductor L is not  0 , the current continues to flow through the internal diode of Q 2 . In this case, because the internal diode has high ON voltage in comparison to the MOS structural part, the conduction loss becomes large compared to the case that the MOS structural part is ON. Therefore, the time resolution of td 2  is occasionally required more than that of td 1  for improving the efficiency. In this case, the number of the delay elements D of the delay circuit  93  is set to be larger than the number of the delay elements D of the delay circuit  92 . 
     At any rate, in the delay circuits  92 ,  93 , it is necessary that the number of the delay elements is set to be smaller than that of the delay circuit  91  and that the supplied current is also set to be smaller than that of the delay circuit  91 . 
     Eleventh Embodiment 
       FIG. 40  is a circuit diagram illustrating a structure of a digital PWM circuit  87  according to an eleventh embodiment of the invention. 
     In the above-described tenth embodiment, as shown in  FIG. 39 , the set pulse D 2 _set of D 2  counts MSB[duty+td 1 ] by the counter  46 , and compares LSB[duty+td 1 ] therewith by the comparator  57 , and inputs the signal to the delay circuit  92 . In this structure, for the dead time td 1 , a negative value can also be set. The negative value of td 1  corresponds to the case that the falling timing of D 2  is set to be before the falling timing of D 1 . When the ON time of the low-side switching element is extremely larger than the OFF time of the high-side switching element, td 1  is required to be set to be negative, but in general, the high-side and low-side switching elements whose ON time and OFF time are extremely different are not used. Accordingly, td 1  rarely becomes negative. 
     Accordingly, in this embodiment, as shown in  FIG. 40 , the signal (output of the multiplexer  101 ) generating the reset pulse of D 1  is input to the delay circuit  92  generating the set pulse D 2 _set of D 2 . Thereby, rising of D 2  is generated on the basis of falling of D 2 , and therefore, the counter  46  and the comparator  57  at the former stage of the delay circuit  92  becomes unnecessary, and the circuit scale can be smaller than that of the tenth embodiment. 
     In this embodiment, in the same manner as the tenth embodiment, in each of the delay circuits  92 ,  93 , the number of the delay elements is set to be smaller and the supplied current is also set to be smaller than that of the delay circuit  91 , and thereby, the time resolution dt 2  of each of the D 2 _set signal determining the rising timing of the driving signal D 2  and the D 2 _reset signal determining the falling timing thereof may be longer than the time resolution dt 1  of the driving signal D 1  (dt 1 &lt;dt 2 ), and also, dt 2 =dt 1  is possible. 
     Moreover, in the same manner as the tenth embodiment, in the delay circuit  92  and the delay circuit  93 , the number of the delay elements is set to be the same m, and the supplied current is also set to be the same Isrc 2 , but the number of the elements or the supplied current may be set to be different between the two delay circuits  92 ,  93 . 
     Twelfth Embodiment 
     Next, one specific example of the structure corresponding to the comparator  3  and the A/D conversion circuit  4  in the structure shown in the above-described  FIG. 14  will be explained.  FIG. 21  is a schematic view illustrating a structure of DC-DC converter according to a twelfth embodiment of the invention. The same signs are appended to the same components as the above-described embodiments. 
     The output voltage Vout of the switching power circuit having the switching elements Q 1 , Q 2  and the inductor L and the condenser C is compared to the target output voltage in the error signal generation circuit having the D/A conversion circuit  7  and the comparators  3   a  to  3   f  and the A/D conversion circuit  4 , and the error signal e corresponding to the difference between the output voltage Vout and the target output voltage is generated. 
     Specifically, the output voltage Vout is input to the comparator  3   a  to  3   f  and compared with the target output voltage (reference voltage Vref). In the D/A conversion circuit  7 , as shown in  FIG. 22 , a plurality of stages (six in the example shown in the figure) of Bin (thresholds) are set, and supplied to each of the comparators  3   a  to  3   f  as the analog voltages. 
     The comparator  3   a  compares Vref and (Vref+q/2), and the comparator  3   b  compares Vref and (Vref−q/2), and the comparator  3   c  compares Vref and (Vref+3q/2), and the comparator  3   d  compares Vref and (Vref−3q/2), and the comparator  3   e  compares Vref and (Vref+5q/2), and the comparator  3   f  compares Vref and (Vref−5q/2). 
     The A/D conversion circuit  4  receives the comparison result of the comparators  3   a  to  3   f , and outputs the control parameter for compensating the disagreement amount of the output voltage Vout with respect to the target output voltage (reference voltage Vref), as the error signal (digital value) e to the compensator  5 . 
     The A/D conversion circuit  4  determines the error signal e with reference to Look Up Table in which the corresponding relation between Bin (q/2, −q/2, 3q/2, −3q/2, 5q/2, −5q/2) and the control parameter (q, −q, 3q, −3q, 5q, −5q) is preliminarily written. 
     When the output voltage Vout becomes larger than Vref and disagrees to reach (Vref+q/2), the error signal e corresponding to the control parameter “q” is output, and when the output voltage Vout becomes smaller than Vref and disagrees to reach (Vref−q/2), the error signal e corresponding to the control parameter “−q” is output, and when the output voltage Vout becomes larger than Vref and disagrees to reach (Vref+3q/2), the error signal e corresponding to the control parameter “2q” is output, and when the output voltage Vout becomes smaller than Vref and disagrees to reach (Vref−3q/2), the error signal e corresponding to the control parameter “−2q” is output, and when the output voltage Vout becomes larger than Vref and disagrees to reach (Vref+5q/2), the error signal e corresponding to the control parameter “3q” is output, and when the output voltage Vout becomes smaller than Vref and disagrees to reach (Vref−5q/2), the error signal e corresponding to the control parameter “−3q” is output. 
     The compensator  5  calculates the duty command value d on the basis of the error signal e and outputs the duty command value d to the digital PWM circuit  6 . For example, with reference to Look Up Table in which the corresponding relation between the error signal e and the duty command value d calculated with respect to the error signal is preliminarily written, the duty command value d is determined. 
     Like the above-described embodiments, the digital PWM circuit  6  generates a switching pulse provided to the gate of the high-side switching element Q 1  and a switching pulse provided to the gate of the low side switching element Q 2 , and supplies the switching pulses to each of the switching elements Q 1 , Q 2 . 
     Thirteenth Embodiment 
     Next, with reference to  FIGS. 24 to 26 , a thirteenth embodiment of the invention will be explained.  FIGS. 24 ,  25 ,  26  correspond to above-described  FIGS. 21 ,  22 ,  23 , respectively. 
     When the output voltage Vout is changed rapidly by rapid change of the charge, if the disagreement amount with respect to Vref comes to exceed the uppermost or lowermost Bin, the control parameter becomes quite the same (3q or −3q) in the region thereof no matter how far Vout is separate from Vref, and therefore, when Vout disagrees so as to be extremely larger than (Vref+5q/2) or when Vout disagrees so as to be extremely smaller than (Vref−5q/2), Vout does not converge to Vref only by the control of one time of the control parameter (3q or −3q), and the control by the control parameter (3q or −3q) becomes repeated at several times, and time is required for making Vout converge to Vref. That is, if Bin width and the control parameter are fixed, the control response with respect to large rapid change of the charge is degraded. 
     Accordingly, in this embodiment, to the uppermost Bin (Vref+5q/2) and the lowermost Bin (Vref−5q/2), parameters of (+a) and (−a) can be added, respectively. Furthermore, as shown in  FIG. 26 , the control parameter (3q+b) corresponding to Bin [(Vref+5q/2)+a] and the control parameter (−3q-b) corresponding to Bin [(Vref−5q/2)−a] are written in Look Up Table. For a, a plurality of values can be set, and correspondingly, b can be a plurality of values. 
     First, for example, a=0 and b=0 are set. When the load rapidly decreases and the output voltage Vout exceeds (Vref+5q/2), the value of a is set to be a larger value (correspondingly the value of b becomes larger), the uppermost Bin is changed from (Vref+5q/2) to [(Vref+5q/2)+a], and at the same time, the control parameter (3q+b) corresponding to Bin[(Vref+5q/2)+a] is read from Look Up Table, and the error signal e corresponding to the control parameter (3q+b) is output. 
     That is, when the output voltage Vout disagrees to be larger than Vref, by enhancing Bin width correspondingly, the larger control parameter can be used and the output voltage Vout can be made to converge to Vref in a short time, and response in the control operation of setting the output voltage Vout to be a desired target value can be enhanced. 
     Also, the case that the load rapidly increases can be thought similarly, and when the output voltage Vout becomes smaller than (Vref−5q/2), the value of a (absolute value) is set to be larger than 0, and the lowermost Bin is changed from (Vref−5q/2) to [(Vref−5q/2)−a], and simultaneously, the control parameter (−3q−b) corresponding to Bin [(Vref−5q/2)−a] is read from Look Up Table, and the error signal e corresponding to control parameter (−3q−b) is output. 
     As described above, according to this embodiment, in operating the power circuit, the uppermost or lowermost Bin width is changed according to the change width of the output voltage Vout, and the control parameter is changed to be a larger control parameter, and thereby, the response with respect to the large rapid change of the charge (convergent property of Vout to the target value) can be improved. 
     In the above-described embodiment, the specific examples of changing uppermost and lowermost Bin widths has been exemplified, but by appropriately changing the Bin widths of the other positions similarly, the improvement of response can be expected. 
     Next, the embodiment that a plurality of DC-DC converters are connected in parallel and driven in parallel will be explained. 
     Fourteenth Embodiment 
       FIG. 27  is a schematic view showing a structure of a digital control power source according to a fourteenth embodiment. 
     A plurality of DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N are connected in parallel between the input voltage source  1  (see,  FIG. 14 ) and the output terminal. The output side of the inductor L of each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N is connected in parallel to the common output line  90 . In the output line  90 , a condenser C and the load  2  are commonly connected to each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N. 
     The power ICs  30 - 1 ,  30 - 2 , . . .  30 -N of the respective DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N correspond to the power IC  30  surrounded by dashed line in  FIG. 14 . As the digital PWM circuit  6  and the comparator  3  and the A/D conversion circuit  4  in each of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N, the ones having the structures described above with reference to  FIGS. 1 to 26  and  38  to  40  can be appropriately used. 
     The power ICs  30 - 1 ,  30 - 2 , . . .  30 -N have the same structures, and the inductors L also have the same structures, and therefore, the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N have the same structures, and the same output voltage can be output in principle. Furthermore, each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N (the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N) has the capability of becoming the master (function of being capable of supplying reference signal or the like to the other converters). 
     When the output current of each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N is, for example, 10 amperes, if, for example, 10 converters are connected in parallel and operated in parallel, a power source having an output current of 100 amperes can be composed. This can be more insufficient than the case that one DC-DC converter having an output current of 100 amperes is used. 
     Moreover, the output voltage of each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N has a ripple as shown in  FIG. 28A , but when each of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N is operated (multiphase-operated) so that phases of output voltages of the respective DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N come to disagree little by little as shown in  FIG. 28B , the ripple can be reduced. 
     The power ICs  30 - 1 ,  30 - 2 , . . .  30 -N are connected to a common (one) data bus  22 . Furthermore, the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N are connected to a common (one) synchronization signal line  21 . Through them, the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N communicate to one another. 
     First, on start-up, the power IC to be a reference of the output voltage phase 0° is determined. Here, for example, the power IC  30 - 1  is made to be the power IC. The power IC outputs the internal reference clock from SYNC terminal (not shown) to the synchronization signal line  21 , and the other power ICs  30 - 2 , . . .  30 -N receive the reference clock from each of the SYNC terminals through the synchronization signal line  21 , and operates with being synchronized with the reference clock by using, for example, the above-described structure such as PLL. That is, all of the DC-DC converters  50 - 1 ,  50 - 2 , . . .  50 -N operate with being synchronized with the same reference clock. 
     Moreover, the power IC  30 - 1  indicates a phase shift value of the output voltage through the data bus  22  to the other power ICs  30 - 2 , . . .  30 -N. Each of the power ICs  30 - 2 , . . .  30 -N except for the power IC  30 - 1  receives the phase shift value through the data bus  22 , and as shown in  FIG. 28B , phases of output voltages of the respective power ICs  30 - 1 ,  30 - 2 , . . .  30 -N are made to disagree with one another, and the ripple is reduced. 
     All of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N operate with being synchronized with the reference clock of the power IC  30 - 1 . Therefore, if a trouble is caused in the reference clock inside the power IC  30 - 1 , the entire system becomes failed. For preventing this, the power IC  30 - 1  has a function of detecting its own trouble, and if the trouble is detected, the sequence data of which power IC is next made to function as the master is indicated through the data bus  22 . Therefore, all of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N can be the reference IC having a phase of 0° and have a function of indicating the phase shift value to the other power ICs. If a trouble is caused in the power IC functioning as the master, another normally operating power IC functions as the master instead of the troubled power IC, and therefore, the reliability of the entire system can be enhanced. 
     As described above, in this embodiment, the power ICs communicate to one another and can autonomously set the interleaving, and therefore, even if a trouble is caused in any one power IC, the system of the entire DC-DC converters is not failed, and therefore, the reliability can be enhanced. 
     Fifteenth Embodiment 
     In a digital control power source in which N DC-DC converters (power ICs) having the same structures are driven in parallel, for maximizing efficiency (output power/input power), how each of the power ICs is operated is important. In particular, when the entire output current comes to decrease in such a case as light load, compared to the case in which the current of the entire current I divided by N (I/N) is flowed to each of N power ICs, the efficiency becomes high in the case in which the operation of some power ICs out of the N power ICs is stopped and only Nop (&lt;N) power ICs are operated and the current of (I/Nop) is flowed to each of the operating power ICs. 
     In general, when the output current becomes small in the DC-DC converter, the efficiency becomes lower as shown in  FIG. 29 . In this case, consumption current of the control circuit or power for charging and discharging the gate of the power stage (switching elements Q 1 , Q 2 ) does not change and does not decrease even when the output current of the DC-DC converter changes. Accordingly, as the output power comes to decrease, the efficiency (output power/input power) lowers. 
     When the entire output current decreases and therewith the output current of each of N power ICs lowers and the efficiency of the current becomes lower than the peak, operation of at least one or more power ICs out of N power ICs is stopped and the current flowing through the individual operating power source ICs is held to be high, and thereby, the efficiency can be held to be high. The power ICs whose operation is stopped are naturally the power ICs except for the power IC functioning as the master. 
     In the example of  FIG. 29 , when the value of (entire current I/operation number Nop) becomes smaller than the current Imax in which the efficiency becomes the peak, if the operation number Nop is reduced to suppress the current value (I/Nop) flowing through each of the power ICs to be in the vicinity of Imax, the efficiency can be held to be high. 
     When Nop power ICs are operating now and the entire output current is I, the current of approximately (I/Nop) flows through each of the power ICs. And, when (I/Nop) becomes smaller than the preliminarily determined current value, X power ICs out of Nop power ICs are stopped. In order that the residual (Nop-X) power ICs share and continuously flow the entire current I, the value of the current flowing through each of the (Nop-X) power ICs is set to be [Nop/(Nop-X)]−times larger. 
     For the power IC whose operation is stopped, because operations of a large part of the control circuit and the power stage (switching elements Q 1 , Q 2 ) are stopped, the power consumption becomes almost zero. The operating circuit comes to be only the part waiting so as to be capable of operating by receiving the input of the signal when the current increases. 
     In determining the number Nop of the power ICs made to operate, as describe later, (I/Nop) by which each of the power ICs maintains the operation by the continuous mode is preferable. 
     In general, a DC-DC converter has operation regions of a continuous mode and discontinuous mode. The continuous mode is an operation mode in which the inductor current flowing to the direction of supplying the current to the load is larger than 0, and the discontinuous mode is an operation mode including a period in which the inductor current flowing to the direction of supplying the current to the load becomes 0. 
     In particular, if the current becomes small in the case of light load, when the high-side switching element Q 1  is OFF and the low-side switching element Q 2  is ON, the inductor current becomes zero or negative current in which the current is not supplied to the load, and the operation becomes the discontinuous mode. In the example shown in  FIG. 29 , when the current (I/Nop) flowing through each of Nop operating power ICs (each of DC-DC converters) becomes I 0  or less, the operation becomes the discontinuous mode. 
     As shown in  FIG. 29 , in the region that (I/Nop) is I 0  or less, the efficiency lowers more and more, and in the region of larger than I 0 , there is the value Imax of (I/Nop) of maximizing the efficiency. Accordingly, by adjusting the number Nop of the operating power ICs so that each of the power ICs always operate in the continuous mode (so that I/Nop becomes larger than I 0 ), lowering of the efficiency is suppressed and the operation in the vicinity of the maximum efficiency becomes possible. 
     Moreover, in particular, in the digital control DC-DC converter, the transfer function in the control system is different between the continuous mode and the discontinuous mode, and therefore, the control is required to be changed, but by setting the operating power ICs to be in a state of the continuous mode, the same transfer function can be used for the control system of each of the power ICs, and the control becomes simple. 
     Sixteenth Embodiment 
     In the DC-DC converters made to operate in parallel, it is desirable that as described above, the output current of each of the DC-DC converters becomes (I/operation number Nop) when the current flowing through the load is I, but by manufacturing variation or the like despite the power ICs the same structures, there are variations of the internal reference voltage (reference voltage Vref and the offset voltage of the A/D conversion circuit  4 , and also, element characteristics such as each of the output FETs (switching elements Q 1 , Q 2 ) or the LC filter vary, and therefore, unless the output voltage control loop is set to be one and positive current balance control is performed, it is difficult to flow the uniform current through each of the DC-DC converters (each of power ICs). 
     When each of the power ICs regulates the output (controls the output to the target voltage) by the control loop using its own output voltage control circuit (the comparator  3 , the A/D conversion circuit  4 , the compensator  5 , and so forth), for example only Vref of the power IC  30 - 1  varies to be higher with respect to the other power ICs  30 - 2 , . . .  30 -N, almost all of the entire current I comes to concentrate on the DC-DC converter  50 - 1 , and the current flowing through the other DC-DC converters  50 - 2 , . . .  50 -N becomes almost zero. 
       FIG. 30  is a schematic view showing a structure of a digital control power source according to a sixteenth embodiment of the invention. The same signs are appended to the same components as above-described  FIG. 27 , and the detailed explanation thereof will be omitted. 
     Each of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N are connected in parallel to a common (one) error share bus  23 . Each of the operating power ICs controls ON/OFF duty of the switching elements Q 1 , Q 2  on the basis of the error signal e obtained in the error signal generation circuit (the comparator  3 , the A/D conversion circuit  4 ) as explained in the above-described embodiment. In this embodiment, the error signal e obtained in any one of the power ICs is shared and used by the operating entire power IC. 
     For example, if the power IC  30 - 1  is made to be the power IC functioning as the master, the power IC  30 - 1  calculates the duty command value d by the compensator  5  by using the error signal e obtained by its own A/D conversion circuit  4  and controls the output voltage on the basis of the duty command value d, and transmits the error signal e through the error share bus  23  to the other operating power ICs. The other power ICs calculate the duty command value d by the respective compensators  5  on the basis of the error signal e, and control ON/OFF of the switching elements Q 1 , Q 2 , and thereby control the output current. 
     That is, because ON/OFF control of the switching elements Q 1 , Q 2  is performed based on the common error signal e in each of the operating power ICs, independent from the characteristic variation of the error signal generation circuit part (the comparator  3 , the A/D conversion circuit  4 , and so forth) among the power ICs, the output current can be evenly distributed among the power ICs, and the current can be prevented from concentrating on one Power IC. 
     Not the error signal e but the duty command value d may be shared and used among the operating entire power ICs. That is, the power IC  30 - 1  calculates the duty command value d by the compensator  5  by using the error signal e obtained by its own A/D conversion circuit  4  and control the output voltage and transmits the duty command value d through the error share bus  23  to the other operating power ICs. The other power ICs control On/OFF of the respective switching elements Q 1 , Q 2  on the basis of the duty command value d. 
     Because ON/OFF control of the switching elements Q 1 , Q 2  is performed based on the common duty command value d in each of the operating power ICs, independent from the characteristic variation of the comparator  3  and the A/D conversion circuit  4  and the compensator  5  and so forth among the power ICs, the output current can be evenly distributed among the power ICs, and the current can be prevented from concentrating on one Power IC. 
     In performing the above-described control, it is necessary to consider the delay time in which the power IC  30 - 1  functioning as the master calculates the voltage compensation information (the error signal e or the duty command vale d) and then transmits the information to the other operating power IC and then the other power ICs drive ON/OFF of the switching elements Q 1 , Q 2  on the basis of the received voltage compensation information. 
     Here, in the multiphase operation that each of the power ICs makes the output phases disagree with one another, the power IC  30 - 1  functioning as the master calculates the above-described voltage compensation information at the times of the number of the operating power ICs (at Nop times) and transmits the voltage compensation information sequentially from smaller phases with respect to the phase 0° of the power IC  30 - 1 , and thereby, the delay time between the timing that the power IC  30 - 1  transmits the voltage compensation information to the target power IC and the timing that the switching elements Q 1 , Q 2  of the power ICs receiving the information start to switch is suppressed to be small, and thereby, the control response to the target output voltage can be enhanced. And, additionally, by the multiphase operation, power system in which the output ripple is suppressed can be realized. 
     Moreover, in the transient response, by using not the voltage compensation information from the power IC  30 - 1  but its own comparator  3  and the A/D conversion circuit  4  and the compensator  5 , the error signal e or the duty command value d is calculated to perform the transient response, and thereby, the response property can be more improved. In this case, in the static state, in the power ICs  30 - 2 , . . .  30 -N except for the power IC  30 - 1 , it is preferable to adjust the internal reference voltage (reference voltage Vref) so that the error signal e that is the same as the error signal e transmitted from the power IC  30 - 1  can be generated. Thereby, because the variation of the reference voltage of each of the power IC can be compensated, the error among the phase currents can be smaller. 
     Seventeenth Embodiment 
     By the above-described sixteenth embodiment, the output current of each of the power ICs is balanced uniformly to some extent, but the characteristic variation of the output FETs (switching elements Q 1 , Q 2 ) or the LC filter of each of the power ICs cannot be compensated, and therefore, higher precise current balance control becomes required. 
       FIG. 31  is a schematic view showing a structure of a digital control power source according to a seventeenth embodiment of the invention. The same signs are appended to the same components as the  FIGS. 27   30 , and the detailed explanation thereof will be omitted. 
     In this embodiment, in addition of the above-described synchronization signal line  21  and the error share bus  23 , to a common (one) current share bus  24 , the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N are connected in parallel. Each of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N transmits and receives the current value data through the current share bus  24  with being synchronized with the clock signal supplied through the synchronization signal line  21 . 
     The power ICs  30 - 1 ,  30 - 2 , . . . ,  30 -N have a driver Dv and a receiver Rv connected to the current share bus  24  as shown in  FIG. 32 . In  FIG. 32 , each of the power ICs  30 - 1 ,  30 - 2 , . . .  30 -N is shown representatively by the sign  30 . 
     To the driver Dv, the signal T of its own current value converted into a pulse width is input.  FIG. 33  shows an example that the current value is converted into the high-level pulse width W 1  with being synchronized with the clock signal X. The signal T is inverted in the driver Dv and output to the current share bus  24 , and also input to its own receiver Rv. 
     The driver is a so-called an open drain type. In this embodiment, the open drain output terminal of the driver Dv of each of the power ICs is connected in parallel to the current share bus  24 , and additionally, pulled up to the power source by the resistance R shown in  FIG. 31 , and the function of the negative-logic wired OR (wired OR) is realized. That is, when the driver Dv of any one of the power ICs is outputting “Low”, the current share bus  24  becomes in the “Low” level. 
     This will be explained by using  FIG. 34  as the example. Each of the power ICs is synchronized with the signal synchronized with the switching cycle and outputs the pulse signals Y- 1 , Y- 2 , . . . Y-n having pulse widths corresponding to the respective current values to the current share bus  24 . When the current value is converted to the high-level pulse width W 1  like the signal T shown in  FIG. 33 , the current value is inverted by the driver Dv and therefore the low-level width corresponds to the current value in each of the signals Y- 1 , Y- 2 , . . . Y-n. 
     In the example shown in  FIG. 34 , because the low-level pulse width of the signal Y- 2  is the longest, the signal Y in the current share bus  24  comes to have the same wave shape as the signal Y- 2  by the negative logic wired OR, and the common signal Y is input to the receiver Rv of each of the power ICs. 
     The signal Y is converted in the receiver Rv to be the signal S, and from the high-level pulse width of this signal S, the maximum current value can be detected in the output current values of the respective operating power ICs. That is, each of the operating power ICs can share the maximum current value data as the share current value data. 
     The minimum current value of the output current values of the operating power ICs may be shared among the power ICs as the share current value data. 
     That is, when each of the operating power ICs converts its own current value into the low-level pulse width W 2  as shown in  FIG. 35 , the signal T is inverted by the driver Dv and therefore the high-level width corresponds to the current value in the each of the signals Y- 1 , Y- 2 , . . . Y-n in  FIG. 34 . 
     And, the signal Y comes to have the same wave shape as the signal having the largest low-level pulse width, and conversely, the signal having the smallest high-level pulse width is shared among the power ICs as the signal Y. In this case, because the current value is converted to the high-level width, it corresponds to the minimum current value that the high-level width is the smallest. 
     In this embodiment, for example, the power ICs are connected in a ring shape to compose a token ring, and the transmission sequence of the current value data is determined on start-up, the power ICs taking in the token (transmission right) are made to sequentially transmit the current value data to the current share bus  24 . 
     The time T required for recognizing the current value data of the entire operating phases can be represented by T=Nphase×Ts/Xphase, in which Nphase is the number of operation phases and Ts is switching period and Xphase is the number of phases of the current value data shared in one switching cycle. 
     By using the share current value data obtained as described above, each the operating power ICs composes the current compensation loop as shown in  FIG. 36  in addition of the above-described voltage compensation loop, and compensates the duty command value d obtained by the voltage compensation loop. 
     That is, each of the power ICs compares the share current value data obtained through current share bus  24  and the own output current value (inductor current value) iL by the comparator  12 , and the comparison result is output to the current compensator  13 , and the current compensator  13  calculates the duty compensation value Δd on the basis of the comparison result. The Δd corresponds to the duty control amount for controlling iL to be the current corresponding to the above-described share current value data. 
     When the current error that each of the power ICs subtracts its own current value from the share current value is ei, the transfer function Gdi(z) from ei to Δd can be exemplified in the following formulas 1-1, 1-2. In this case, b 0  to b 2  are constants. 
     
       
         
           
             
               
                 
                   
                     
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     When the transfer function Gdi(z) is known, by an inverse z transformation of transforming the transfer function (frequency axis) to a difference equation, the output (Δd) can be calculated from the input (ei). In the case of Gdi(z) of the above-described formulas 1-1, 1-2, Δd becomes the following formulas 2-1, 2-2, respectively. [n] represents the number n of the sample data. In the above-described treatment, after the A/D conversion treatment, the signal is discretized.
 
Δ d[n]=Δd[n− 1 ]+b   0   e[n]   2-1
 
Δ d[n]=Δd[n− 1 ]+b   0   e[n]+b   1   e[n− 1 ]+b   2   e[n− 2]  2-2
 
     Δd calculated by the current compensator  13  is biased in the accumulator  18  with respect to the duty command value d calculated in the voltage compensator  5 , and (d+Δd) is supplied to the digital PWM circuit  6  as the duty command value, if only the current compensation loop is viewed, the control is that the current is uniformed to the maximum current value or the minimum current value among the power ICs, but each of the power ICs individually performs the duty control so that the voltage becomes the target voltage (reference voltage Vref) by the above-described voltage compensation loop, and therefore, by performing the switching control of the switching elements Q 1 , Q 2  on the basis of (d+Δd), the output current of each of the power ICs is compensated to the direction to becoming smaller (larger) when the current is too large (too small), and therewith, the current is uniformed among the power ICs. 
     That is, each of the power ICs shares the voltage compensation information (the error signal e or the duty command value d) through the error share bus  23 , and thereby the current nonuniformity due to characteristic variation of the compensator  3  and the A/D conversion circuit  4  and the voltage compensator  5  and so forth can be compensated, and additionally, the share current value data is shared through the current share bus  24 , and thereby, the current nonuniformity due to characteristic variation of the output FETs (switching elements Q 1 , Q 2 ) or LC filter can be compensated. As a result, the current balance control among the power ICs can be performed higher-precisely and the current concentration on one power IC can be avoided. 
     It is not necessary that the current share bus  24  and the error share bus  23  are separated as the hardware, but a common bus can also be used by sectioning the time so that the voltage compensation information is communicated in one time period of the inside of the switching cycle and that the share current value data is communicated in the other time period. 
     In this embodiment, in sharing the share current value data required for the control to evenly uniform the output current among a plurality of DC-DC converters (power ICs) connected in parallel, high-speed digital bus is not used, and the power consumption is small. Moreover, the reference voltage Vref is not shared among the power ICs, and the increase of the number of the terminals can be suppressed, and furthermore, all of the communications are performed by the digital method, and therefore, are difficult to be affected by the wiring delay. 
     The abnormal state (error) of the power system includes input overvoltage, input low-voltage, input overcurrent, input overpower, output overvoltage, output low-voltage, output overcurrent, output overpower, output tracking error, overtemperature, low-temperature, nonvolatile memory read error, and so forth. 
     In recent years, market requirement for power management is enhanced, and it is necessary that the control (power-system protection) methods in the various power system abnormalities can be finely set for each of the abnormal states. 
     On the other hand, for various errors, the power system can be protected by controlling the output-stage FET. For example, the case that the output overcurrent error and the output overvoltage error are simultaneously caused is thought. When the power IC is set so that the output current is maintained to be constant in the output overcurrent error and so that the output is stopped in the output overvoltage, because the controllable part is only the output-stage FET, the operation simultaneously satisfying the above-described settings cannot be realized and contradiction is caused. 
     Accordingly, for avoiding this, in the embodiments of this invention, the priority sequence is determined for each of errors in each of the above-described power systems, and when a plurality of errors are caused at the same time, the errors are addressed from higher priority sequence. The priority sequence can be set in the sequence of higher risk for the power system, and can also be programmable through a communication bus or a nonvolatile memory. 
     As described above, the embodiments of this invention have been explained with reference to the specific examples. However, this invention is not limited thereto, and various modifications based on the technical ideas of this invention are possible. 
     In the above-described embodiments, the voltage step down DC-DC converter has been exemplified as the digital control power, but this invention is not limited thereto and is applicable to voltage step up DC-DC converter or other voltage conversion circuits. Moreover, the digital PWM circuit of this invention illustrated in  FIGS. 1 to 20  is applicable to other applications such as motor driver and LED (Light Emitting Diode). 
     According to a fourth aspect of the invention, there is provided a semiconductor device comprising: a switching power circuit having a switching element; an error signal generation circuit for detecting an output voltage of the switching power circuit and comparing the output voltage with a target voltage and generating a error signal on the basis of the comparison result thereof; and a digital PWM circuit for supplying a pulse signal whose duty is determined based on the error signal to compensate disagreement of the output voltage with respect to the target voltage, to a control terminal of the switching element, the error signal generation circuit outputting a control parameter determined based on a corresponding relation between a threshold set to have a plurality of stages according to disagreement width with respect to the target voltage and the control parameter set with according to the threshold, a width of the threshold being changed in operating the switching power circuit, and the control parameter changing with corresponding to the change of the width of the threshold. 
     There is provided a device according to the fourth aspect, wherein when the output voltage becomes larger than the uppermost threshold, the width of the uppermost threshold is changed to be larger. 
     There is provided a device according to the fourth aspect, wherein when the output voltage becomes smaller than the lowermost threshold, the width of the lowermost threshold is changed to be smaller. 
     There is provided a device according to the fourth aspect, further comprising a compensator calculating the duty on the basis of the error signal. 
     According to a fifth aspect of the invention, there is provided a semiconductor device comprising a plurality of digital control power sources having same structures connected in parallel with respect to a common output line, a current flowing each of the operating digital control power sources out of the plurality of digital control power sources being controlled so that each of the operating digital control power sources maintains operation in a continuous mode. 
     According to a sixth aspect of the invention, there is provided a semiconductor device comprising: a plurality of digital control power sources having same structures connected in parallel with respect to a common output line; and a signal line connected in parallel so that the plurality of digital control power source can communicate with one another, each of the digital control power sources having an output voltage control circuit controlling each of output voltages to be a target voltage, voltage compensation information obtained by the output voltage control circuit of any one of the operating digital control power sources out of the plurality of digital control power sources being shared among all of the operating digital control power sources through the signal line, the operating digital control power sources controlling the output voltage on the basis of the shared voltage compensation information. 
     There is provided a device according to the sixth aspect, wherein the output voltage control circuit has an error signal generation circuit for detecting the output voltage and comparing the output voltage with the target voltage and generating an error signal corresponding to the disagree amount of the output voltage with respect to the target voltage, and each of the digital control power sources sharing the error signal. 
     There is provided a device according to the sixth aspect, wherein each of the digital control power sources has a switching element, and the output voltage control circuit has: an error signal generation circuit for detecting the output voltage and comparing the output voltage with the target voltage and generating an error signal corresponding to the disagree amount of the output voltage with respect to the target voltage; and a compensator for calculating duty of pulse signal switching ON/OFF of the switching element on the basis of the error signal, and each of the digital control power sources sharing the duty. 
     According to a seventh aspect of the invention, there is provided a semiconductor device comprising: a plurality of digital control power sources having same structures connected in parallel with respect to a common output line; and a signal line connected in parallel so that the plurality of digital control power source can communicate with one another, each of the operating digital control power sources outputting current value data of its own output current value converted into a pulse width to the signal line and recognizing the current value data corresponding to the maximum current value or the minimum current value from the current value data of all of the operating digital control power sources as share current value data, and each of the operating digital control power sources controlling the output current on the basis of the share current value data. 
     There is provided a device according to the seventh aspect, wherein each of the digital control power sources has an open drain output terminal, and the open drain output terminals are connected in parallel to compose wired OR. 
     There is provided a device according to the seventh aspect, wherein phases of output voltages of digital control power sources disagree with one another. 
     There is provided a device according to the seventh aspect, wherein any one of the plurality of digital control power sources indicates phase shift values of the respective output voltages to the other digital control power sources. 
     According to a eighth aspect of the invention, there is provided a semiconductor device comprising a digital PWM circuit outputting: a first pulse signal switching from a low level to a high level at a timing of being synchronized with a clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal; and a second pulse signal switching from a low level to a high level at a timing set by a shorter time resolution than that of a period of the clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal, the digital PWM circuit including: a first delay circuit having a plurality of stages of first delay elements connected serially; a first selection circuit selecting one from outputs of the respective delay elements and outputting a signal setting a falling timing of the first pulse signal; a second delay circuit having a plurality of stages of second delay elements connected serially; a second selection circuit selecting one from outputs of the respective second delay elements and outputting a signal setting a rising timing of the second pulse signal; a third delay circuit having a plurality of stages of third delay elements connected serially; and a third selection circuit selecting one from outputs of the respective third delay elements and outputting a signal setting a falling timing of the second pulse signal, a delay amount in the second delay elements being longer than a delay amount in the first delay elements, and a delay amount in the third delay elements being longer than a delay amount in the first delay elements. 
     There is provided a device according to the eighth aspect, wherein a control current supplied to the second delay elements is smaller than a control current supplied to the first delay elements, and a control current supplied to the second delay elements is smaller than a control current supplied to the third delay elements. 
     There is provided a device according to the eighth aspect, wherein the number of stages of second delay elements is smaller than the number of stages of first delay elements, and the number of stages of third delay elements is smaller than the number of stages of first delay elements, 
     There is provided a device according to the eighth aspect, wherein an output of the first selection circuit is input to the first-stage second delay element in the second delay circuit. 
     According to a ninth aspect of the invention, there is provided a semiconductor device comprising a digital PWM circuit outputting: a first pulse signal switching from a low level to a high level at a timing of being synchronized with a clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal; and a second pulse signal switching from a low level to a high level at a timing set by a shorter time resolution than that of a period of the clock signal and switching from the high level to the low level at a timing set by a shorter time resolution than that of a period of the clock signal, the digital PWM circuit including: a first delay circuit having a plurality of stages of first delay elements connected serially; a first selection circuit selecting one from outputs of the respective delay elements and outputting a signal setting a falling timing of the first pulse signal; a second delay circuit having a plurality of stages of second delay elements connected serially; a second selection circuit selecting one from outputs of the respective second delay elements and outputting a signal setting a rising timing of the second pulse signal; a third delay circuit having a plurality of stages of third delay elements connected serially; and a third selection circuit selecting one from outputs of the respective third delay elements and outputting a signal setting a falling timing of the second pulse signal; an output of the first selection circuit being input to the first-stage second delay element in the second delay circuit. 
     There is provided a device according to the ninth aspect, wherein the first delay elements, the second delay elements, and the third delay elements have same structures.