Patent Publication Number: US-2006013412-A1

Title: Method and system for reduction of noise in microphone signals

Description:
BACKGROUND OF THE INVENTION  
      1. Field of the Invention  
      This invention relates generally to the fields of directional microphones, microphone arrays, noise reduction and sound enhancement.  
      2. Description of the Related Art  
      Widely used omnidirectional microphones pick up sounds (including various interferences) coming from different directions equally well. It means that noise, echoes, room reverberation and other interferences can significantly degrade quality of signals recorded by such microphones. In order to improve a signal-to-noise ratio (that is a ratio between levels of a useful signal and interfering signals picked up by a microphone), a wide range of means for reduction of noise in microphone signals has been developed.  
      One of the simplest and widely used approaches to improve the signal-to-noise ratio (SNR) for microphone signals is represented by directional microphones. The directional microphones attenuate sounds coming from particular directions, so that in case interferences are coming from directions, different from that of the signal of interest, they can be attenuated, with a proportionate SNR improvement.  
      By far the most popular type of the directional microphone is a first order gradient type microphone. This type of microphone can be designed employing acoustic or electronic means.  FIG. 1  shows a general scheme of a simple prior art electronic directional microphone generally indicated as  5 PA. The directional microphone  5 PA comprises two omnidirectional microphones  10  spaced apart by a distance d along a microphone axis A-A coincident with an expected direction of a sound wave S 0 (t) carrying a useful signal. The microphone  10 F receiving first the sound wave S 0 (t) is usually termed a front microphone, while the other microphone  10 R is termed a rear microphone. The distance d between the microphones  10  has to be large enough to provide a detectable phase shift for low frequencies. At the same time, said distance must be less than half of the shortest acoustic wavelength in the operative frequency range to avoid spatial aliasing (see Clarkson, Peter M., Optimal and Adaptive Signal Processing, CRC Press, 1993). This corresponds to usable distances between the front microphone  10 F and the rear microphone  10 R in the range of 10 to 40 mm.  
      The prior art electronic directional microphone further includes: a delay line  12  receiving an output signal R(t) of the rear microphone  10 R and producing a signal R(t−τ) (delayed in respect to the signal R(t) by a preset time delay τ); and a subtracter-adder  14  for subtracting the signal R(t−τ) from an output signal F(t) of the front microphone  10 F.  
      In a particular case of a sound wave S(t) of unit amplitude and frequency ƒ forming an angle Θ with the microphone axis A-A, an output of such directional microphone is given by the following equation:
 
 D (ƒ,Θ)= e   −j2πƒt (1 −e   −j2πƒ(τ+T cos(Θ)) ),  (1)
 
 where Θ, τ, ƒ are as specified above, T=d/V sound  is a sound propagation time between the microphones  10 F,  10 R, and V sound  is the sound velocity. Taking the magnitude of Eq. 1 yields
 
| D (ƒ,Θ)|=2|sin(πƒ(τ+ T  cos(Θ)))|  (2)
 
      Assuming a relatively small distance between the microphones and a small delay (ƒd/V sound &lt;&lt;1 and ƒτ&lt;1), we obtain:
 
| D (ƒ,Θ)|= 2 πƒ(τ+   T  cos(Θ))  (3)
 
      Varying the delay τ between 0 and T, it is possible to get, using the directional microphone  5 PA shown in  FIG. 1 , different polar patterns. For example, τ=0 leads to a bi-directional microphone, τ=T leads to a cardioid pattern, τ=0.5T leads to a super-cardioid pattern.  
      One can see from Eq. 3 that, by varying τ between 0 and T, it is possible to steer the null in the back plane between 90° and 270°. The null cannot be moved to the front plane, and thus, the signal coming from front directions with θ between −90° and 90° cannot be canceled.  
      In principle, it is possible to make the delay τ adjustable. The choice of an optimal delay τ opt  depends on the acoustic conditions such as the room reverberation as well as a number, spectral content and a direction of interfering signals. If an appropriate digital signal processor (DSP) is used to perform the delay and subtract operations, then τ may be adjusted automatically to provide the best directivity according to some criteria. However, even with this addition, noise reduction effectiveness of the simple directional microphone of  FIG. 1  is limited in many important respects.  
      For example, in case when interferences have different spectral contents, the optimal delay τ opt  can be different for different frequency bands. Thus, in this case uniform delay in the whole frequency range does not allow to achieve the maximal possible SNR improvement. Further, it follows from Eq. 3 that different values of τ correspond to different front (θ=0) sensitivities inside 6 dB range. Such difference must be automatically compensated to provide constant frequency response.  
      Eq. 3 shows also that for a fixed τ sensitivity is proportional to the frequency. If a flat frequency response is required, such proportionality should be compensated accordingly. For a small distance between the microphones (where Eq. 3 is valid) such compensation can be achieved by multiplying the output in the frequency domain by the factor 1/ƒ. A problem with such normalization arises, when short time RMS values of sound pressure levels on the two microphones  10  are not equal. This happens for example when the distance between the microphones  10  and the sound source becomes comparable to the distance d between the microphones, so that a “far field” assumption is not valid. A resulting excessive low frequencies amplification due to multiplication by 1/ƒ is called “a proximity effect”. Another example of insufficiency of the described normalization is wind turbulences, when short time RMS values of sound pressure levels on the two microphones fluctuate independently.  
      A further problem arises in cases of a mismatch between sensitivities of two microphones serving as parts of the described directional microphone. For all such cases the normalized output is given as
 
 D   q (ƒ,Θ)= e   −j2πƒt (1− qe   −j2πƒ(τ+T cos(Θ)) )/ƒ,  (4)
 
 where the value of q indicates the degree of the mismatch. Division by ƒ corresponds to a normalization that is necessary to provide a flat frequency response corresponding to an ideal case (q=1). For zero delay τ=0 and the front sound direction Θ=0 the output amplitude is given by
 
| D   q (ƒ)|=|1− qe   −j2πƒT |/ƒ.
 
 The frequency response of such microphone relative to the ideal one (q=1) is correspondingly given as  
                 B   q     ⁡     (   f   )       =                D   q     ⁡     (   f   )                   D   ⁡     (   f   )              =            1   -     q   ⁢           ⁢     ⅇ       -   j2π     ⁢           ⁢   fT                       1   -     ⅇ       -   j2π     ⁢           ⁢   fT                          (   5   )             
 
 Eq. 5 shows that, depending on the mismatch q, there may be a significant excessive amplification of low frequencies. For example, for the distance between the microphones equal to 15 mm and relatively small mismatches q=0.9 (expressed in decibels 20 log10(0.9)≅−1 dB mismatch), and q=0.8 (20 log10(0.8)≅−2 dB mismatch)
 
 B   q=0.9  (100 Hz)≅3.7≅11.5 dB
 
 B   q=0.8  (100 HZ)≅7.3≅17.3 dB
 
      To avoid or to alleviate the described and other disadvantages and/or limitations of the simple directional microphone system, many more elaborate methods and systems for processing electrical signals derived from omnidirectional microphones have been designed. U.S. Pat. No. 4,653,102 discloses a system for reduction of noise in microphone signals in a far talk mode by employing two directional microphones and a microcomputer for performing a fast Fourier transform of received signals in order to go from the time domain to the frequency domain, said transform being followed with an area and phase sorting aimed at improving SNR for a wanted sound in a well-defined area, and with an inverse fast Fourier transform. Use of the Fourier transform and the inverse Fourier transform in combination with a manipulation of frequency domain data to produce a noise-reduced signal is described also in U.S. Pat. No.6,668,062.  
      U.S. Pat. No. 5,182,774 discloses a headset supplied with an earcup and means for generating the anti-noise signal from the microphone signal obtained from a directional microphone, which detects and transduces the acoustical pressure within the earcup cavity. Another headset design that utilizes active noise cancellation and a booster circuit to compensate for low frequency losses when active noise cancellation is in operation is presented in U.S. Pat. No. 5,604,813.  
      The system described in U.S. Pat. No. 5,664,021 uses a combination of two directional microphones, mixing circuitry, and control circuitry to simulate a signal that would be generated by a single directional microphone pivoted to direct its maximum response at the acoustic signal as the acoustic signal moves about the environment. According to U.S. Pat. No. 6,584,203A, tracking a moving noise source can be performed with an aid of a second-order adaptive differential microphone array (ADMA).A subband implementation of the ADMA can be used for tracking a different moving noise source for each different frequency subband.  
      A dual microphone noise reduction system intended for use in mobile phones and employing a far-mouth microphone in conjunction with a near-mouth microphone is disclosed in U.S. Pat. No. 6,549,586. Speech enhancement is attained by including spectral subtraction algorithms using linear convolution, causal filtering and/or spectrum dependent exponential averaging of the spectral subtraction gain function.  
      U.S. Pat. No. 5,917,921 describes a noise reducing microphone apparatus having a pair of microphone units and an adaptive noise canceller receiving a primary input from one of the microphone units and a reference input from another microphone unit. In the adaptive noise canceller, the reference input is subtracted from the primary input through an adaptive filter, which adaptive filter is adaptively controlled by an output signal resulted from the subtraction in such a way as to minimize an output power of the system.  
      Notwithstanding a substantial progress in regard to noise reduction achieved in modem microphone systems through an application of various methods of digital signal processing, a long-felt need still exists for versatile and cost-effective microphone systems capable to provide sufficient noise reduction and sound enhancement of microphone signals in various far-talk and/or close-talk applications.  
     BRIEF SUMMARY OF THE INVENTION  
      Accordingly, the main object of this invention is to provide a method and a system for reduction of noise in microphone signals, said method and system of the invention possessing the following advantageous features: 
          a) an improved noise canceling when the system is used in any of the close and far talk modes;     b) an automatic compensation for different frequency response of the microphones;     c) a reduced sensitivity to wind turbulence;     d) an automatic compensation or control of the proximity effect;     e) minimal distortions of the signal of interest irrespective to the level of the noise or interfering sound.        

      It is another object of the present invention to provide a compact microphone system suitable for mobile applications.  
      It is a further object of the invention to provide a directional microphone system demanding only relatively simple digital signal processing of input signals suitable for implementation on relatively inexpensive digital signal processors with fixed point arithmetic.  
      These and other objects of the present invention are achieved primarily by employing a selective approach to digital processing of microphone signals depending on a particular operative mode of the noise reduction system of the present invention, with the main feature of said selective approach consisting in using a specific constraint on digital filtering of one of microphone signals for each of two main operative modes. More precisely, it was found that, when the system of the invention functions in the far talk operative mode, the optimal form of the said constraint consists in making any of the filter coefficients nonnegative. On the other hand, the optimal form of the said constraint when using the close talk operative mode corresponds to limiting a sum of absolute values of the filter coefficients not to exceed a predetermined value.  
      A basic method implementing the described selective approach and corresponding to the first aspect of the present invention comprises the following main steps: 
          (a) providing a front digital signal and a rear digital signal by converting to digital form electrical signals from a front microphone and a rear microphone;     (b) producing a filtered rear signal by filtering the rear digital signal through an application thereto of continuously adaptable filter coefficients;     (c) producing a subtracted signal by subtracting the filtered rear signal from the front digital signal; and     (d) continuously adapting said filter coefficients by supplying the rear digital signal and the subtracted signal to adapting means, said adapting means configured to keep any of the filter coefficients nonnegative, when functioning in the far talk operative mode, and/or to restrict the sum of absolute values of the filter coefficients not to exceed a predetermined value, when functioning in the close talk operative mode.        

      According to a preferred embodiment of the invention, the method of noise reduction in microphone signals further comprises a step (e) of optionally performing additional processing of the subtracted signal. When the far talk mode is employed. such processing preferably comprises: 
          computing, on the base of the filter coefficients used in step (b), an equalization coefficient;     producing an equalized signal by multiplying the subtracted signal by the equalization coefficient;     computing, on the base of the front digital signal, the rear digital signal and the equalized signal, a scaling coefficient; and     producing a processed signal by multiplying the equalized signal by the scaling coefficient.        

      According to another preferred embodiment of the method of the invention, each of the digital signals produced on the base of the front and rear microphone signals is split into M frequency subband signals, and steps (b), (c), (d) and (e) are performed in parallel for each group of signals corresponding to one of the subbands. Then all processed subband signals are combined to form a processed noise reduced signal.  
      In its second aspect the invention provides a system for implementing the described noise-reduction method.  
      In its simplest version, the system of the present invention comprises a digital signal processor having at least one adaptive processing unit. The or each adaptive processing unit comprises at least: 
          input terminals for receiving a front digital signal and a rear digital signal; and     an adaptive filtering unit comprising: 
            filter means for filtering the rear digital signal through an application thereto of continuously adaptable filter coefficients;     subtracting means for subtracting a filtered rear signal from the front digital signal; and     adapting means for receiving the rear digital signal and the subtracted signal; for continuously adapting said filter coefficients; and for supplying the adapted filter coefficients to the filtering means.    
               

      The adapting means is advantageously configured to keep any of the filter coefficients nonnegative, when functioning in the far talk operative mode, and/or to restrict the sum of absolute values of the filter coefficients not to exceed a predetermined value, when functioning in the close talk operative mode. The purpose of the constraints employed in each operative mode is to preserve the signal of interest while reducing the interfering signals.  
      When adapted to implement any or each of the preferred embodiments of the inventive method, the system of the invention can further comprise, in appropriate combinations, such parts, as: 
          a front microphone and a rear microphone producing a front microphone signal and a rear microphone signal;     a front input channel and a rear input channel, said channels configured to receive the front microphone signal and the rear microphone signal and to convert them into a front input digital signal and into a rear input digital signal;     a band equalizer block configured to receive the filter coefficients from the adaptive filtering unit and to compute, on the base of said filter coefficients, an equalization coefficient;     first multiplication means for producing an equalized signal by multiplying the subtracted signal by the equalization coefficient;     an output level controller configured to receive the front digital signal, the rear digital signal and the equalized signal and to compute, on the base of said signals, a scaling coefficient; and     second multiplication means for producing a processed signal by multiplying the equalized signal by the scaling coefficient.        

      The preferred embodiments of the invention perform several additional functions, including a normalization of the output signal to compensate reduced sensitivity for low frequencies; turbulence noise reduction; and proximity effect control.  
      In case the method of the invention includes the steps of splitting digital signals obtained from the front and rear microphone signals into M frequency subband signals and parallel processing each group of signals corresponding to one of the subbands, the digital signal processor of the noise-reduction system further comprises M adaptive processing units; a first splitter for splitting the front input digital signal into M frequency subband signals and a second splitter for splitting the rear input digital signal into M frequency subband signals. Also, the system further comprises means configured for receiving the processed signal from each adaptive processing unit and for combining said processed signals into a processed noise reduced signal.  
      To adapt the system of the invention for functioning selectively either in the far talk or close talk operative mode, the system is preferably provided with a mode selector configured for selectively generating either a far talk mode selecting signal or a close talk mode selecting signal, wherein the adapting means or each adapting means is further adapted for receiving the selecting signal to trigger the adapting means into a far talk operative mode or a close talk operative mode.  
      By extending the concept of the present invention from two to a larger number of omnidirectional microphones, second order directivity in the far talk mode can be achieved. In other words, the system of the present invention can be implemented as an autodirective quadruple microphone comprising two pairs of omnidirectional microphones. The adaptive processing unit (or each of M adaptive processing units, in case the above described splitting into M subbands is provided) of such autodirective quadruple microphone is structured into a first processing block and a second processing block. While the second processing block by its structure and functions is similar to the described adaptive processing unit of the basic embodiment of the system, the first processing block may be described as comprising an adaptive filtering unit consisting of two filter blocks and two subtracter-adders, but only one adaptive coefficients block. This means that said adaptive coefficients block receives signals from both filter blocks and, in its turn, supplies filter blocks with filter coefficients identical for both filter blocks.  
      The above-described and further objects, features and advantages of the present invention will become apparent from the following detailed description of the preferred embodiments taken in conjunction with the following drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       FIG. 1  shows a general scheme of a prior art electronic directional microphone system;  
       FIG. 2  shows a general scheme of the first embodiment of the present invention utilizing two microphones;  
       FIG. 3  is a block diagram of applying means and a digital signal processor for the system shown in  FIG. 2 ;  
       FIG. 4  is a block diagram of an adaptive processing unit for the digital signal processor of  FIG. 3 ;  
       FIG. 5  is a block diagram of an output level controller for the adaptive processing unit of  FIG. 4 ;  
       FIG. 6  shows a general scheme of the second embodiment of the present invention utilizing four microphones;  
       FIG. 7  is a simplified block diagram of a digital signal processor for the system shown in  FIG. 6 , and  
       FIG. 8  is a block diagram of a processing block for the digital signal processor of  FIG. 6 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION  
       FIG. 2  shows the general scheme of the first embodiment of the noise reduction system according to the invention. The system of the invention is implemented as an improved autodirective dual microphone system of the general type shown in  FIG. 1 . Similarly to the prior art system, the proposed autodirective dual microphone system comprises two spatially distant microphones  10 , a front microphone  10 F and a rear microphone  10 R.  
      The front microphone  10 F and the rear microphone  10 R are connected correspondingly to a front input channel and a rear input channel, each of said channels being represented by an analog-to-digital converter (ADC)  20 F,  20 R. When the microphones  10 F,  10 R receive acoustic signals, they correspondingly produce, in response to sound pressure changes, a front microphone signal F(t) and a rear microphone signal R(t), said signals F(t), R(t) being continuous analog electric signals. On receiving signals F(t), R(t), the analog-to-digital converters  20 F,  20 R of the input channels transform them into front and rear digital signals F(n), R(n).  
      In their turn, the front and rear input channels are connected to applying means (schematically represented in  FIG. 2  as two arrows). Said applying means supply the input digital signals R(n), F(n) to a digital signal processor (DSP) generally designated as  30 . The DSP  30  can be implemented on a special digital signal processor, a general-purpose processor, an Application Specific Integrated Circuit (ASIC) and/or by other appropriate digital means.  
       FIG. 3  shows the schematic of a preferred embodiment of the DSP  30 . The DSP comprises a first splitter  50 F connected to the first input channel (not shown) for receiving therefrom the front digital signal F(n) and a second splitter  50 R connected to the second input channel (not shown) for receiving therefrom the rear digital signal R(n). The splitters  50 R,  50 F split each of the rear and front input digital signals R(n), F(n) into M frequency subband signals, namely into front and rear subband signals {F b (n)}, {R b (n)}. As shall be evident to persons skilled in the art, a proper designed digital IIR or FIR filter bank can be used for implementing the splitters  50 R,  50 F. Alternatively, FFT based subband decomposition can be used.  
      As schematically shown in  FIG. 3 , the DSP  30  comprises M adaptive processing units (APU) numbered as  60   1  . . .  60   m . All adaptive processing units have a similar or identical design and one of them, the APU  60   b , is illustrated as a block diagram in  FIG. 4 . As can be seen from  FIG. 4 , each APU comprises a first terminal  62  for receiving corresponding front subband signal F b (n) and a second terminal  64  for receiving corresponding rear subband signal R b (n). In other words, as shown in  FIG. 3 , the first terminal and the second terminal of the first APU  60 , receive the first front subband signal F 1 (n) from the first splitter  50 F and the first rear subband signal R 1 (n) from the second splitter  50 R, while the first terminal  62   b  and the second terminal  64   b  of the bth APU  60   b  (see  FIG. 4 ) receive correspondingly the bth front subband signal F b (n) from the first splitter  50 F and the bth rear subband signal R b (n) from the second splitter  50 R. Thus, pairs {R b (n),F b (n)}, that is pairs of a front digital signal and a rear digital signal of corresponding subband signals constitute an input of each of M adaptive processing units  60 .  
      As will be described in detail below, each APU  60  produces optimal directivity signal P b (n) in the frequency subband allotted to said APU. Output subband signals {P b (n)} may be further processed by an optional processor  70  (schematically represented in  FIG. 3 ) before they are combined into the full band noise reduced signal A(n) by a combiner  80 . The optional processor  70  is adapted to perform different digital signal processing tasks that are generally performed in frequency subbands. Some of such tasks will be mentioned below.  
      As can be further seen from  FIG. 4 , according to a preferred embodiment of the invention, the first input terminal  62  and the second input terminal  64  of the represented APU  60   b  are correspondingly connected to a first upsampling block  130   1  and to a second upsampling block  130   2 , so that the front and the rear digital signals F b (n), R b (n) constitute input signals for said first and second upsampling blocks. As will be explained below, upsampling of the front and rear input signals is necessary to provide sufficient time resolution between signal samples.  
      The main part of each APU  60  is constituted by an adaptive filtering unit  85 , said unit providing all the directivity and noise canceling functionality. The adaptive filtering unit  85  consists of filter means formed as a filter block  90 ; subtracting means formed as a subtracter-adder  92 ; and adaptive means formed as an adaptive coefficients block  95 . Both the filter block  90  and the adaptive coefficients block  95  are connected to the second input terminal  64  via the second upsampling unit  130   2  for receiving an upsampled rear digital signal, while one of entrances of the subtracter-adder  92  is connected to an exit of the filter block  90  to receive a filtered rear signal therefrom.  
      According to the preferred embodiment, the proposed noise reduction system of the invention is configured for functioning either in a far talk operative mode or in a close talk operative mode. In the far talk mode the interfering signals are considered to be all signals coming from the rear hemisphere relative to the microphone axis. For all such signals the front microphone signal is delayed relative to the rear microphone signal. In the close talk mode the interfering signals are considered to be all signals that are relatively far away from the microphone. For all such signals the ratio of amplitudes of the front and rear microphone signals are close to unity.  
      Switching between said operative modes is performed by means of a mode selector  35  (see  FIG. 3 ) adapted to generate a control signal C. The control signal C is generated at either one of two levels, a first of said levels (i.e. high or low level C F ) corresponding to the far talk mode selecting signal, and a second one (i.e. low or high level C C ) corresponding to the close talk mode selecting signal. As shown in  FIG. 4 , the control signal C is applied to the adaptive coefficients block  95 , the band equalizer  110  and the output level controller  120 .  
      The mode selector  35  also controls, by applying the control signal C, a mode switch  100  that connects, either directly or via a delay line  105 , the first upsampling block  130   1  to the second entrance of the subtracter-adder  92 . As shown, the delay line  105  is enabled in the close talk mode and bypassed in the far talk mode.  
      A band equalizer  110  is supplied with an output signal A b (n) from the adaptive coefficients block  95 . An equalization coefficient q b (n) from the exit of the band equalizer  110  is applied to one of entrances of first multiplication means formed as a first multiplicator  115 . Another entrance of the multiplicator  115  is connected to the exit of the subtracter-adder  92 . The connection between the subtracter-adder  92  and the first multiplicator  115  is made via a downsampling block  140 .  
      An equalized signal Q b (n) from the first multiplicator  115  is applied to one of the entrances of an output level controller  120 , which controller serves to prevent possible excessive output signal amplification. One of the entrances of the output level controller  120  is connected to the mode selector  35  for receiving therefrom the control signal C. Two remaining entrances of the output level controller  120  are connected to the first and the second input terminals  62 ,  64  for receiving the first and the second digital input signals.  
      A preferred structure of the output level controller  120  is shown in  FIG. 5 . The output level controller comprises three similar blocks  122 , each said blocks being adapted for receiving one of the rear input signal R b (n), the front input signal F b (n) or the equalized signal Q b (n) and for determining a level of a received signal.  
      The output level controller  120  comprises also a scaling coefficient calculator  124  performing the computation of the scaling coefficient r b (n), as will be described in more detail below. As can be seen from  FIG. 4 , the scaling coefficient is applied to one of the entrances of a second multiplicator  125 , with another of its entrances being connected to the exit of the multiplicator  115  (see  FIG. 4 ) for receiving therefrom the equalized signal Q b (n). The processed signal P b (n) produced by the second multiplicator  125  is applied to the output terminal  66  of the APU  60   b.    
      The band equalizer  110 , the output level controller  120  and two multiplicators  115 ,  125  constitute a preferred embodiment of processing means of each of the APU  60 .  
      The processed signals P b (n) from each of the APU  60  are fed into a combiner  80  that produces a full band digital output signal P(n) (see  FIG. 3 ). If additional processing of the processed subbband signals P b (n) is desirable, it can be accomplished by an optional processor  70 . Examples of such processing include but not limited to noise suppression, multiband signal compression, time-scale speech modification, echo canceling, etc.  
      Functioning of the APU  60   b  in the far talk and close talk operative modes according to a preferred version of the method of the invention will be now described with reference to  FIG. 4 .  
      First, the functioning of the APU in the far talk operative mode will be considered.  
      Far Talk Operative Mode  
      As explained above, the simultaneous switching of all APU  60  into the far talk mode is performed by a generation by the mode selector  35  of the control signal C at the first preset level C F  corresponding to this mode. Setting the level of the control signal to the C F  can be performed by an operator of the system of the invention by means of a corresponding switch or button (not shown) provided in the mode selector  35  or by any other appropriate means, i.e. from a keyboard, from some distant control system, etc. The generation of the C F  signal also results in switching on the output level controller  120  and in bypassing of the delay line  105  (that is in connecting the upsampling block  130   1  directly to the subtracter-adder  92  of the adaptive filtering unit).  
      In the far talk mode the delay line  105  is bypassed and the adaptive filter length is set as
 
 L=N+ 1,  (6)
 
 where N is proportional to the sound propagation time between the microphones  10 . N can be calculated as
 
 N=[d R   s   /V   sound ],  ( 7) 
 
 where d is the distance between the microphones, R s  is the sampling rate of analog-to-digital converters  20 , V sound  is the sound velocity in the air and [] denotes here the operation of truncation to the nearest integer value. 
 
      As can be seen from  FIG. 4 , the front and rear digital signals F b (n), R b (n) supplied correspondingly to the first and second input terminals  62 ,  64  are first upsampled in the first and second upsampling blocks  130   1 ,  130   2 . The expediency of the upsampling step is explained by that, in order to use lower computational resources, it is preferable to work with the lowest possible sampling rate R s  of analog-to-digital converters  20  in the input channels. As explained, for example, in Proakis, John G., Digital Signal Processing. Principle, Algorithms and Applications, Prentice-Hall, 1996, for correct operation without spectral aliasing, sampling rate R s  must exceed twice the highest frequency in a signal being digitized. The typical sampling rate used in digital voice communication is 8 kHz, which means that the upper frequency in all analog signals before analog-to-digital conversion must be limited to 4 kHz. However, with such low sampling rates R s  the time resolution provided by adjacent samples can be insufficient for the present invention to operate effectively. Indeed, for 8 kHz sampling rate and the distance between the microphones  10  equal to 35 mm, Eq. 7 gives:
 
 N≅[ 0.035·8000/341]≅[0.8]=0
 
      According to Eq. 6, N=0 corresponds to unit length L of adaptive filter ( 90 ), and, hence, no directivity options except a bi-directional microphone are possible in this case. That is why, in order to provide variable directivity options, sampling rates of the digital input signals R b (n), F b (n) first should be increased in upsampling blocks 130 by a factor K to provide better time resolution between samples. For example, for the distance between the microphones  10  equal to 35 mm and upsampling factor K equal to 4
 
 N≅[ 0.035·4·8000/341]≅[3.3]=3.
 
      This provides enough resolution for most of applications. As shown in the above-cited book of Proakis, upsampling may be accomplished by inserting K zeros between every original sample and filtering the result with a corresponding low-pass digital filter.  
      The upsampled rear input digital signal is supplied to the filter block  90  of the adaptive filtering unit  85 . The filter block  90  filters said rear input digital signal by applying thereto filter coefficients, which are calculated by the adaptive coefficients block  95 . The purpose of adaptive filtering is to remove, to a possible degree, interfering signals from the front microphone signal. According to the present invention, specific constraints are imposed on the coefficients of the filter block  90  to guarantee preservation of the main signal coming from the front direction.  
      As mentioned above, the kind of applied constraints and specific features of some other steps of the method of the invention are determined by the selected operative mode of the noise reduction system.  
      When the system of the invention functions in the far talk mode, with each new sample n of the input digital signal the adaptive filtering unit  85  performs the following sequence of operations: 
          1. Computes an estimate {tilde over (F)} b (n) of F b (n) from the last L samples of R b (n) as:  
                   F   ~     b     ⁡     (   n   )       =       ∑     k   =   0       L   -   1       ⁢         W     b   ,   k       ⁡     (   n   )       ⁢         R   b     ⁡     (     n   -   k     )       .                 (   8   )             
    2. Computes an output sample as the estimation error:
 
 A   b ( n )= F   b ( n )− {tilde over (F)}   b ( n ).  (9)
    3. Updates the filter coefficients to reduce the average output power E{A b   2 (n)} with the following constraint: all filter coefficients are nonnegative.        

      In the formulae above an index b corresponds to the bth frequency subband.  
      Step 1 of filtering is performed by the filter block  90 ; Step 2 of subtracting the filtered rear signal from the front digital signal F b (n) is performed by the subtracter-adder  92 ; and Step 3 of adapting (by updating) the filter coefficients is performed by the adaptive coefficients block  95 . In the preferred embodiment of the present invention Step 3 is performed using a kind of Normalized Least Mean Squares (NLMS) algorithm (see the above-cited Clarkson book) as:  
                   W     b   ,   k       ⁡     (   n   )       =         W     b   ,   k       ⁡     (     n   -   1     )       +         α   ⁢           ⁢       A   b     ⁡     (   n   )             μ   b     ⁡     (   n   )         ⁢       R   b     ⁡     (     n   -   k     )             ,     
     ⁢     k   =       0   ⁢           ⁢   …   ⁢           ⁢   L     -   1               (   10   )             
 
 where μ b (n) is a normalization factor depending on the amplitude of the signal and α is so-called adaptation constant that defines the trade-off between adaptation speed, stability and filter coefficient error in the presence of noise. In the classical NLMS algorithm the normalization factor μ b (n) is computed as  
                 μ   b     ⁡     (   n   )       =       ∑     k   =   0       L   -   1       ⁢       R   b   2     ⁡     (     n   -   k     )                 (   11   )             
 
      In the preferred embodiment of the present invention the normalization factor μ b (n) is computed as
 
μ b ( n )= L ·max(γ·μ b ( n− 1), R   b ( n )) 2 , 0&lt;γ&lt;1  (12)
 
      Such normalization factor works like a peak detector, where γ defines how fast the peak value is forgotten. Similar to Eq. 11, μ b (n) computed according to Eq. 12 reduces the adaptation step when the signal is strong. However, it reacts faster and it is easier to compute.  
      In the preferred embodiment of the present invention the following constraint is imposed on filter coefficients W b,k  in the far talk mode: all filter coefficients are forced to be nonnegative after every filter update:
 
 W   b,k ( n )=max( W   b,k ( n ), 0),  k= 0  . . . L− 1  (13)
 
      The output sample computed according to Eq. 8 represents a subtracted signal A b (n), which signal is supplied from the adaptive filtering unit  85  to the downsampling block  140 . Also, as shown in  FIG. 4 , the same signal A b (n) is supplied to the adaptive coefficients block  95  to enable the computation of the filter coefficients according to Eq. (14).  
      In the far talk mode the subtracted signal A b (n) corresponds to an output signal from a directional microphone of a differential type with directivity pattern changing according to current conditions. According to Eq. 3 for small distances between the microphones  10 , an amplitude of such signal grows linearly with frequency. Therefore the output of such directional microphone must be equalized to provide a flat frequency response for far sounds coming from the front directions with Θ=0. According to the method of the present invention, such equalization is performed by multiplying the subtracted signal of every filter block  90  by dynamically changing equalization coefficient depending on the current filter coefficients. The equalization coefficient q b (n) is supplied by the band equalizer  110 . In the preferred embodiment of the invention said coefficient is computed as:  
                   q   b     ⁡     (   n   )       =     1          1   -       ∑     k   =   0       L   -   1       ⁢           W   _       b   ,   k       ⁡     (   n   )       ·     ⅇ       -   j2π     ⁢           ⁢         f   _     b     ⁡     (       Δτ   ·   k     +     d   /     V   sound         )                          ,           (   15   )             
 
 where coefficients of bth filter W b,k  are normalized to sum to 1 as  
             W   _       b   ,   k       ⁡     (   n   )       =       W     b   ,   k           ∑     i   =   0       L   -   1       ⁢       W     b   ,   i       ⁡     (   n   )               
 
 and {overscore (ƒ)} b  is the central frequency of bth frequency subband. In the preferred embodiment {overscore (ƒ)} b  is computed as.  
             f   _     b     =         f   b   +     +     f   b   -       2       ,       
 
 where ƒ b   + , ƒ b   −  are respectively upper and low cutoff frequencies of the corresponding bandpass filter used in the splitters  50 . Detailed mathematical substantiation for computing the equalization coefficient according to Eq. 15 is given in Appendix. 
 
      As was already mentioned, the filter coefficients used in the computation of the equalization coefficient q b (n) are supplied to the band equalizer  110  from the adaptive coefficients block  95  of the adaptive filtering unit  85 .  
      Equalization coefficient q b (n) is calculated in the far field assumption of equal sound pressure level on both microphones  10 . If it is not the case, multiplication by q b (n) can lead to excessive output signal amplification. For example, relatively small distance between the microphone and the sound source (e.g. mouth) can lead to significantly larger sound pressure on the front microphone. Air turbulences caused by wind can be another reason for random sound pressured level differences on the microphones. The prevention of a possible excessive output signal amplification caused by different levels of sound pressure on microphones  10  when working in the far talk mode is ensured according to the invention with the aid of the output level controller  120 , which becomes active on receiving the appropriate control signal C F .  
      The excessive amplification is eliminated by restricting the level of processed signal P b (n) at the output terminal  66  of the APU  60  to be not greater than the maximal level of the largest of the raw front and the rear input signals F b (n), R b (n) supplied to the corresponding blocks  122  of the band equalizer  120  (see  FIGS. 4, 5 ). In the preferred embodiment of the present invention the restriction is effected by multiplying (by means of the second multiplicator  125 ) the equalized signal Q b (n) by a scaling coefficient r b (n) computed in the scaling coefficient calculator  124  as
 
 r   b ( n )=√{square root over (min(1, max( L   F,b ( n ), L   R,b ( n )/ L   Q,b ( n )))},  (16)
 
 where L F,b (n),L B,b (n),L Q,b (n) are corresponding instantaneous levels of the digital input signals and the equalized output signal, said levels being determined by corresponding blocks  122  and supplied to the scaling coefficient calculator  124 . 
 
      While a signal level may be defined in different ways, when a preferred embodiment of the blocks  122  is employed, a level of a signal X(n) is calculated as
 
 L   X ( n )=max (β· L   X ( n− 1), | X ( n )|),
 
 where the coefficient β&lt;1 depends on the sampling rate and is chosen so that L X  “forgets” 90% of its peak value in about 5 ms. 
 
      The processed signal P b (n) from the second multiplicator  125  is supplied to the output terminal  66  of the APU  60   b.    
      A digital signal P(n) produced as the output of the DSP  30  may be further transformed into analog signal P(t) by a digital-to-analog converter  40  ( FIG. 2 ). The signal P(t) then can be used as an output A(t) of a standard microphone. Alternatively, the signal P(n) may be used in digital form for further processing.  
      Close Talk Operative Mode  
      The system is switched into the close talk mode by the generation, by the mode selector  35 , of the control signal C at the second preset level C C  corresponding to this mode. Such switching results in enabling of the delay line  105  and disabling of the band equalizer  110  and the output level controller  120 .  
      Because in the close talk mode some parts of the APU  60  (i.e. upsapmling and downsampling blocks  130 ,  140 , the subtracter-adder  92 , etc.) perform their functions in a way identical to that in the far talk mode, only specific features of the close talk mode will be described in detail below.  
      Filter block  85  functions essentially in the same regime; however, due to enabling of the delay line  105  corresponding to N samples, the length L of the adaptive filter increases to
 
 L= 2 N+ 1  (17)
 
      Another specific feature consists in a change of the constraint imposed by the adaptive coefficients block  95 . For the close talk mode the sum of absolute values of the filter coefficients shall not exceed a predetermined value. In other words, said sum is limited after every filter update to some value U max &gt;1:  
               U   =       ∑     i   =   0       L   -   1       ⁢            W   k     ⁡     (   i   )                ⁢     
     ⁢         W   k     ⁡     (   i   )       =         W   k     ⁡     (   i   )       ·     min   ⁡     (     1   ,       U   max     /   U       )                   (   18   )             
 
      In the preferred embodiment of the present invention value U max  is set between 1.5 and 3.  
      Further, because the band equalizer  110  and the output level controller  120  are disabled, no equalization coefficient q b (n) and scaling coefficient r b (n) are generated, so that the processed signal P b (n) supplied to the output terminal  66  of the APU  60   b  is the same as the signal A b (n).  
      The detailed mathematical substantiation of the computational scheme according to the present invention (as specified by Eq. 8-18) is supplied in Appendix.  
      According to the above-described embodiment of the present invention directivity is achieved by subtracting a filtered version of the rear digital input signal R b (n) representing the rear microphone signal from the front digital input signal F b (n) representing the front microphone signal. This, first order directivity corresponds to the first derivative of the sound pressure along the microphone axis. In some applications (related mainly to the far talk mode) first order directivity does not provide enough improvement of signal-to-noise ratio, so that second order directivity would be desirable. Such second order directivity can be achieved according to the principles of the present invention by combining outputs of two first order directional microphones (either conventional microphones or ones designed according to the present invention). However, this solution requires accurate matching of phase characteristics of the directional microphones, which matching is, as a rule, difficult to achieve.  
      More advantageous way to achieve the second order directivity in the far talk mode consists in an appropriate extension of the concept of the present invention from two to a larger number of microphones. An embodiment of the proposed system implemented as an autodirective quadruple microphone generally indicated as  5 Q is presented in  FIG. 6 . The inventive quadruple microphone  5 Q comprises two pairs of omnidirectional microphones. In other words, in addition to the first microphone pair consisting of the front microphone  10 F and the rear microphone  10 R, the system presented in  FIG. 6  comprises an additional microphone pair consisting of an additional front microphone  15 F and an additional rear microphone  15 R. Microphones inside each pair are separated by a distance d 1 , while the microphone pairs are separated by a distance d 2 . In a general case, the distance d 1  can differ from the distance d 2 .  
      Similarly to the previously described noise-reduction system of  FIG. 2 , in the system of  FIG. 6  the front microphone  10 F and the rear microphone  10 R are connected correspondingly to the front input channel and the rear input channel, each of said channels being represented by a corresponding analog-to-digital converter  20 . In the same way, the additional front microphone  15 F and the additional rear microphone  15 R are connected correspondingly to the additional front input channel and the additional rear input channel, each of said channels being represented by a corresponding additional analog-to-digital converter  22 . All front and rear input digital signals F 1 (n), R 1 (n), F 2 (n), R 2 (n) produced by the analog-to-digital converters  20 ,  22  on receiving analog electric signals F 1 (t), R 1 (t), F 2 (t), R 2 (t) are applied via applying means to a digital signal processor (DSP)  30 .  
       FIG. 7  shows the schematic of a preferred embodiment of the digital signal processor  30  corresponding to the noise reduction system of  FIG. 6  implementing the autodirective quadruple microphone system. Similar to the DSP of the autodirective dual microphone system shown in  FIG. 3 , the DSP shown in  FIG. 7  comprises two splitters  50 F 1 ,  50 R 1  and  50 F 2 ,  50 R 2  for each microphone pair  10 ,  15 . The front and rear digital signals F 1 (n), R 1 (n), F 2 (n), R 2 (n) are splitted by the splitters  50  into M frequency subband signals, namely into front and rear subband signals {F 1   b (n)}, {R 1   b (n)} and into additional front and rear subband signals {F 2   b (n)}, {R 2   b (n)}.  
      As schematically shown in  FIG. 7 , the DSP  30  in this embodiment of the proposed system comprises M second order adaptive processing units (APU)  150  having identical design and the combiner  80  similar or identical to that shown in  FIG. 3 . Optional processor  70  is omitted in  FIG. 7  for simplicity, while the mode selector  35  generating the control signal C is not used for the reason that this embodiment is intended only for use in the far talk operative mode.  
      Each APU  150  of the autodirective quadruple microphone system differs from the APU  60  in that the APU  150  consists of a first adaptive processing block (APB 1 )  170  and a second adaptive processing block (APB 2 )  180 . Each adaptive processing block  170 ,  180  operates according to the method of the present invention and corresponds to one (a first or a second) stage of processing digital signals representing 4 microphone signals F 1 (t), R 1 (t,) F 2 (t), R 2 (t). The first processing block  170  is fed with two pairs of subband signals and at its output it generates two signals F 3 , R 3 , said signals corresponding to two first order autodirective dual microphone signals with matching phase characteristics. Consequently, said two signals are fed into the second processing block  180  producing an output signal corresponding to a second order directional microphone signal.  
       FIG. 8  shows the schematic of the first processing block  170   b  belonging to the APU  150   b . As will become clear from the following description, the block  170   b  functions as two parallel APU  61   1 ,  61   2  (similar to the above-described APU  60 ) with shared adaptive filter coefficients.  
      The APU  61   1  of the first processing block  170   b  comprises: 
          the first and second terminals (not shown) for receiving correspondingly the front subband signal F 1   b (n) and the rear subband signal R 1   b (n);     the adaptive filtering unit formed by the filter block  90 ; an adaptive coefficients block  200  and the subtracter-adder  92 ;     the processing means comprising the band equalizer  110 , the output level controller  120 , the first multiplicator  115  and the second multiplicator  125 .        

      Similarly, the second of said two APUs, APU  61   2 , of the first processing block  170   b  comprises: 
          the additional first and second terminals (not shown) for receiving correspondingly the additional front subband signal F 2   b (n) and the additional rear subband signal R 2 b(n);     the adaptive filtering unit formed by the filter block  90 ; the adaptive coefficients block  200  and the subtracter-adder  92 ;     the processing means comprising the band equalizer  110 , the output level controller  120   2  the first multiplicator  115  and the second multiplicator  125 .        

      With an exception of the band equalizer  110  and the adaptive coefficients block  200  (which will be discussed in more detail below), all parts of the first and second APU  61   1 ,  61   2  of the first processing block  170   b  are equivalent or identical in their design and functions to the correspondent parts of the of the APU  60   b  described above with reference to  FIG. 5 . Therefore, there is no need in their detailed description. It is sufficient to note that the first and second APU  61   1 ,  61   2  produce at their output terminals correspondingly a first processed signal P 1   b (n) and a second processed signal P 2   b (n) in a way generally similar to the production of the processed signal P b (n) described above with reference to  FIG. 4 .  
      It may be also noted that, though not shown in  FIG. 8 , each of the APU  61   1 ,  61   2  can further comprise two upsampling blocks  130  and one downsampling block  140 , all said blocks having design, connections and functions similar or identical to those of the corresponding blocks of the above-described APU  60   b .  
      As shown in  FIG. 8 , the band equalizer  110  and the adaptive coefficients block  200  are shared by both APU  61   1 ,  61   2 . This means that the adaptive coefficients block  200  receives both the front subband signal F 1   b (n), the rear subband signal R 1   b (n) and a subtracted signal A 1   b (n) from the first APU  61   1  and the additional front subband signal F 2   b (n), the additional rear subband signal R 2 b(n) and a subtracted signal A 2   b (n) from the second APU  61   2 . Operation of the common band equalizer  110  is similar to that of the above-described band equalizer  110 , the only difference being that in the first processing block  170  the band equalizer  110  supplies with the equalization coefficient q b (n) first multiplicators  115  of both APU  61   1 ,  61   2 .  
      The first processing block  150  may be alternatively viewed as comprising an adaptive filtering unit  190  consisting of two filter blocks  90 ,  90 , two subtracter-adders  92 ,  92 , but only one adaptive coefficients block  200 . With each new sample n, adaptive filtering unit  190  performs the following sequence of four operations: 
          1. Computes estimates {tilde over (F)} 1   b (n), {tilde over (F)} 2   b (n) of F 1   b (n), F 2   b (n) from the last L samples of R 1   b (n), R 2   b (n) as:  
                   F   ~     ⁢     1   b     ⁢     (   n   )       =       ∑     k   =   0       L   -   1       ⁢         W     b   ,   k       ⁡     (   n   )       ⁢       R1   b     ⁡     (     n   -   k     )             ⁢     
     ⁢         F   ~     ⁢     2   b     ⁢     (   n   )       =       ∑     k   =   0       L   -   1       ⁢         W     b   ,   k       ⁡     (   n   )       ⁢       R2   b     ⁡     (     n   -   k     )                     (   19   )             
    2. Computes output samples as the estimation errors:
 
 A   1   b ( n )= F   1   b ( n )− {tilde over (F)}   1   b ( n )  (20)
 
 A   2   b ( n )= F   2   b ( n )− {tilde over (F)}   2   b ( n )
    3. Computes filter update normalization constant as:
 
μ b ( n )= L ·max (γ·μ b ( n− 1),  R   1   b ( n ),  R   2   b ( n )) 2 
 
0&lt;γ&lt;1
    4. Updates filter coefficients  
                       W     b   ,   k       ⁡     (   n   )       =         W     b   ,   k       (     n   -   1     )     +       α       μ   b     ⁡     (   n   )         ⁢     (           A1   b     ⁡     (   n   )       ·       R1   b     ⁡     (     n   -   k     )         +         A2   b     ⁡     (   n   )       ·       R2   b     ⁡     (     n   -   k     )           )                     k   =       0   ⁢           ⁢   …   ⁢           ⁢   L     -   1                   (   21   )             
       

      It is evident from the above expressions that general principles of operation of the adaptive filtering unit  190  are equivalent to those of the adaptive filtering unit  85  in the far talk operative mode. However, computations of the estimates and the output samples are conducted independently for signals received from each of the APU  60   1 , APU  60   2 , while computations of the normalization constant μ b (n) and the filter coefficients W b,k (n) are conducted for data sets including signals received from both APU  60   1 , APU  60   2 . Correspondingly, Eq. 21 is equivalent to updating the filter coefficients twice for every n.  
      Using the same filter coefficients for both filter blocks  90  corresponding to the first and second microphone pairs ensures the same phase characteristics of the processed signals. This is important as both the first processed signal P 1   b (n) and the second processed signal P 2   b (n) constituting an output of the first processing block  170  are used as input signals for the second processing block  180 .  
      The second processing block  180  of the adaptive filtering unit  160  is equivalent to the APU  60  (shown in  FIG. 4 ) functioning in the far talk mode. For that reason its schematic is almost the same as presented in  FIG. 4 , but no control signal C and the delay (obtained with the delay line  105 ) are used. As shown in  FIG. 7 , the processed signals outputted from the first processing block  170  are correspondingly applied to input terminals of the second processing block  180  as a front digital signal F 3   b (n) and a rear digital signal R 3   b (n) (said signals correspond to signals F 3   1 (n), R 3   1 (n) for the first APU  150 , and to signals F 3   m (n), R 3   m (n) for the last, mth APU  150   m  shown in  FIG. 8 ). On receiving the pair of signals from the corresponding first processing block  170   b , each of the second processing blocks  180   b  produces, in the way described above in relation to the APU  60   b , an output signal corresponding to a processed subband signal P b (n).  
      Then all subband signals are combined in the described manner inside the combiner  80  into a full band processed signal P(n).  
      When the distances d 1  between microphones  10 ,  15  in each microphone pair and the distance d 2  between the first and second microphone pairs are equal (d 1 =d 2 ), two middle microphones (the rear microphone  10 R of the first pair and the additional front microphone  15 F of the second, additional pair) coincide in space, so it is possible to eliminate one of them (i.e. the additional front microphone  15 F). In this case the microphone signal from the remaining middle microphone  10 R is used both as the rear microphone signal R 1 (t) of the first microphone pair and as the additional front microphone signal F 2 (t) of the additional microphone pair.  
      The invented microphone system using two microphones and a digital signal processor offers the following advantages over conventional prior art microphone systems: 
          improved noise rejection due to fast adaptation of its directive characteristics to the current acoustic conditions;     no on-axis sound distortion;     no “proximity effect” associated with amplification of low frequencies by a standard directional microphone when the distance between the microphone and the sound source becomes comparable to the microphone dimensions;     low sensitivity to wind turbulence noises;     improved noise attenuation comparing to a prior art directive microphone when it is used in a close talk mode;     low implementation cost in terms of computational power and memory consumption;     relatively easy integration into different form factors, mobile or other devices or objects of interior;     low requirements for microphone frequency responses to match each other;     extensibility for building directional microphone systems of higher order.        

      As for the autodirective quadruple microphone system of the present invention, while conserving all important advantages of the dual microphone system, it can provide much better rejection of interfering sounds.  
      This description uses several examples to disclose the invention, including its best mode, and also to enable a person skilled in the art to make and use the invention. It will be obvious to those of ordinary skill in the art that various changes and modifications can be made without departing from the spirit and scope of the invention. The patentable scope of the invention is therefore defined by the claims, and it includes other examples that occur to those skilled in the art.  
      For example, in case requirements for quality of the output processed signal can be made less strict, a step of controlling of the output level of each equalized subband signal Q b (n) can be omitted, with a corresponding simplification of the system of the invention by omitting all output level controllers  120  and all multiplicators  115  associated with said controllers.  
      Further simplification can be attained by omitting all band equalizers  110  and all associated first multiplicators  115 .  
      Even more substantial simplification (again in cases when a tradeoff between processing quality and costs is permissible) can be achieved by omitting the step of splitting each of the input signals into M subband signals. When said splitting step is not used, the system of the invention can be designed without splitters  50  and the combiner  80 . Moreover, only a single adaptive processing unit will be needed.  
      Further, the method and the system of the invention can be implemented, with all above-listed advantages, not only for processing microphone signals in real time, but also when working in an “off-line” regime, that is when microphone or similar signals are recorded in an appropriate recording medium or memory means. If this is the case, then the steps of receiving and converting the microphone signals are not always necessary for implementing the method of the invention. Correspondingly, the microphones themselves do not constitute necessary parts of the inventive noise-reduction system.  
      Further, in cases when the input signals are recorded in a digital form, there is no need to supply the system of the invention with any analog-to-digital converters constituting input channels.  
      All discussed options of simplifying the method and the system of the invention are fully applicable to all described embodiments of the invention, including those corresponding to the autodirective quadruple microphone system presented in FIGS.  6  to  8 . Still further simplification is possible in relation to the embodiment corresponding to the autodirective dual microphone system presented in FIGS.  2  to  5 . More specifically, instead of using such system selectively in either far talk or close talk modes, it can be adapted for only any one of such operative modes. Evidently, such system will not need the mode selector  35  and the mode switch  100 . Moreover, a system intended only for the far talk mode will have no use for the delay line  105 , while a system intended only for the close talk mode will not use the band equalizer  1   10  and the output level controller  120 .  
     APPENDIX  
      Far Talk Mode  
       FIG. 4  shows that in the far talk mode the delay line  105  provided for delaying the front signal F k (n) is bypassed. Eq. 6 states that the maximal delay that may be introduced by the adaptive filter ( 90 ) is equal to L=N samples where N corresponds to the time propagation between the microphones ( 10 ) so that d≅V sound ·N/R s . According to Eq. 8, 9 the output amplitude A for a plane wave of frequency ƒ coming from the direction with angle Θ is given as  
                       A   b     ⁡     (     f   ,   Θ     )       =            F   b     -       R   b     ⁢           ⁢       ∑     i   =   0       L   -   1       ⁢           W   _     b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   ⁡     (       Δ   ⁢           ⁢     τ   ·   i       +     T   ⁡     (   Θ   )         )                                      T   ⁡     (   Θ   )       =     d   ⁢           ⁢   cos   ⁢           ⁢       (   Θ   )     /     V   sound                       Δ   ⁢           ⁢   τ     =     1   /     R   s                     (   22   )               
      The corresponding gain is thus given as  
                 g   b     ⁡     (     f   ,   Θ     )       =         A   b     ⁡     (     f   ,   Θ     )       /     F   b                           ⁢     =          1   -         R   b       F   b       ⁢       ∑     i   =   0       L   -   1       ⁢           W   _     b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   ⁡     (       Δ   ⁢           ⁢     τ   ·   i       +     T   ⁡     (   Θ   )         )                                              ⁢     =          1   -       ∑     i   =   0       L   -   1       ⁢         W   b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   ⁡     (       Δ   ⁢           ⁢     τ   ·   i       +     T   ⁡     (   Θ   )         )                                        W   b     ⁡     (   i   )       =         R   b       F   b       ⁢           ⁢           W   _     b     ⁡     (   i   )       .                 
 
      For 90°≦Θ≦270° the sound propagation delay between microphones corresponds to T(Θ)≦0. It follows that g b (ƒ, Θ)=0 when i=−T(Θ))/ Δτ and W b (i)=R b /F b . Thus, for sounds originating from the back plane a perfect cancellation is achieved. For a mixture of signals coming from directions with 90°≦Θ≦270° a combination of non-negative W b (i) selected such that  
           ∑     i   =   0       L   -   1       ⁢       W   b     ⁡     (   i   )         =       R   b       F   b           
 
 will provide the perfect cancellation. Alternatively, for −90°&lt;Θ&lt;90° and W b (i) restricted to be non-negative, the sound wave is attenuated, but cannot be completely cancelled. For a wave of frequency ƒ coming from front direction (Θ=0) the gain is given by  
           g   b     ⁡     (     f   ,   0     )       =            1   -       ∑     i   =   0       L   -   1       ⁢         W   b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   (       Δ   ⁢           ⁢     τ   ·   i       +     d   /     V   sound         )                    .         
 
      For a plane incident wave (far field case) and equal sensitivities of microphones  10 F,  10 R R b =F b  so that  
                       g   b     ⁡     (     f   ,   0     )       =          1   -       ∑     i   =   0       L   -   1       ⁢           W   _     b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   ⁡     (       Δ   ⁢           ⁢     τ   ·   i       +     d   /     V   sound         )                                and                 ∑     i   =   0       L   -   1       ⁢         W   _     b     ⁡     (   i   )         =   1.                 (   23   )             
 
      To provide a frequency response equal for all frequencies of signals with Θ=0, the output for the narrow band signal with frequency ƒ must be multiplied by the equalization coefficient q b (ƒ) that is inverse to the gain (23)  
         q   b     =       1       g   b     ⁡     (     f   ,   0     )         .         
 
      For a wide band signal each frequency is to be normalized differently. Assuming that the gain difference for frequencies inside each band is small enough, the equalization coefficient q b (n) is computed for the band central frequency {overscore (ƒ)} b  as  
                 q   b     ⁡     (   n   )       =       1       g   b     ⁡     (         f   _     b     ,   0     )         =       1          1   -       ∑     k   =   0       L   -   1       ⁢           W   _       b   ,   k       ⁡     (   n   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢         f   _     b     ⁡     (       Δ   ⁢           ⁢     τ   ·   k       +     d   /     V   sound         )                        .               (   24   )             
 
      Close Talk Mode  
      In the close talk mode there is no preferred direction. All sounds originating outside a close proximity to the microphone are to be cancelled. Positive delays in Eq. 22 make it possible to cancel sounds arriving from directions [90°, 270°]. To cancel sounds arriving from directions [0°, 90°], computations according to Eq. 22 are modified to include negative delays as follows:  
                       A   b     ⁡     (     f   ,   Θ     )       =            F   b     -       R   b     ⁢           ⁢       ∑     i   =   0       L   -   1       ⁢         W   b     ⁡     (   i   )       ·     ⅇ       -   j     ⁢           ⁢   2   ⁢           ⁢   π   ⁢           ⁢     f   ⁡     (       Δ   ⁢           ⁢     τ   ·     (     i   -   N     )         +     T   ⁡     (   Θ   )         )                                    L   =       2   ⁢   N     +   1                   (   25   )             
 
      Introducing a negative delay into the rear microphone signal R b (n) is equivalent to introducing an equivalent positive delay into the front microphone signal F b (n). R b (n). According to the present invention, in the close talk mode the delay line  105  is enabled and the length L of the filter block  90  is computed according to Eq. 17 to incorporate N negative, zero and N positive delays.  
      For a plane incident wave (distant sounds, far field case) and equal sensitivities of microphones  10 F,  10 R R b =F b . With real microphones  
                 γ   ⁢           ⁢     R   b       ≤     F   b     ≤       1   γ     ⁢     R   b         ,           (   26   )             
 
 where γ&lt;1 defines a maximal sensitivity difference between microphones  10 . With good quality microphones γ&gt;0,8 (2 dB). According to the inverse law, the sound pressure amplitude is inversely proportional to the distance to a sound source. Therefore, for sounds generated at zero angle and with ideal microphones  10 F,  10 R:  
                 F   b     =           D   +   d     D     ⁢     R   b       =       B   D     ·     R   b           ,             B   D     =         D   +   d     D     ≥   1               
 
 for all frequency bands, where D is the distance between the sound source and the front microphone  10 F, d is the distance between microphones  100 F,  10 R. For real microphones  
           γ   ·     B   D     ·     R   b       ≤     F   b     ≤       1   γ     ·     B   D     ·     R   b         ,       
 
 where γ&lt;1 again defines the maximal sensitivity difference between microphones  10 . Coefficients W b (i) of the adaptive filter in Eq. 25 are chosen to provide the minimal output signal amplitude. Due to incorporating delays corresponding to sounds coming from all directions, unconstrained filter coefficients W b (i) may provide complete cancellation of all sounds. Amplitude differences caused by factors B and γ are compensated by scaling the filter coefficients accordingly. This is not the desirable situation as close signals with factor B exceeding some threshold must be preserved. This is achieved by constraining the sum of absolute values of coefficients W b (i). After every filter adaptation step the filter coefficients W b (i) are modified as  
       U   =       ∑     i   =   0       L   -   1       ⁢            W   b     ⁡     (   i   )                  
           W   b     ⁡     (   i   )       =         W   b     ⁡     (   i   )       ·     min   ⁡     (     1   ,       U   max     /   U       )             
 
 to satisfy the constraints.