Patent Publication Number: US-8971178-B1

Title: Calibration correction for implicit beamformer using an explicit beamforming technique in a wireless MIMO communication system

Description:
RELATED APPLICATION 
     This is a regular-filed application which claims the benefit of U.S. Provisional Patent Application No. 61/321,046, entitled “Implicit TxBF Calibration With Explicit TxBF,” which was filed on Apr. 5, 2010, the entire disclosure of which is hereby incorporated by reference herein. 
    
    
     FIELD OF TECHNOLOGY 
     The present invention relates generally to wireless communication systems and, more particularly, to a system and method for beamforming while transmitting information in a wireless communication system with multiple transmit antennas and multiple receive antennas. 
     DESCRIPTION OF THE RELATED ART 
     An ever-increasing number of relatively cheap, low power wireless data communication services, networks and devices have been made available over the past number of years, promising near wire speed transmission and reliability. Various wireless technologies are described in detail in the 802.11 IEEE Standard, including for example, the IEEE Standard 802.11 (1999) and its updates and amendments, the IEEE Standard 802.11a/g (2003), as well as the IEEE Standard 802.11n now adopted, all of which are collectively incorporated herein fully by reference. These standards have been or are in the process of being commercialized with the promise of 54 Mbps (802.11a/g) up to 600 Mbps (802.11n) effective bandwidth, making them a strong competitor to traditional wired Ethernet and the more ubiquitous “802.11b” or “WiFi” 11 Mbps mobile wireless transmission standard. 
     Generally speaking, transmission systems compliant with the IEEE 802.11a and 802.11g or “802.11a/g” as well as the 802.11n standards achieve their high data transmission rates using Orthogonal Frequency Division Modulation or OFDM encoded symbols mapped up to a 64 quadrature amplitude modulation (QAM) multi-carrier constellation. In a general sense, the use of OFDM divides the overall system bandwidth into a number of frequency sub-bands or channels, with each frequency sub-band being associated with a respective sub-carrier upon which data may be modulated. Thus, each frequency sub-band of the OFDM system may be viewed as an independent transmission channel within which to send data, thereby increasing the overall throughput or transmission rate of the communication system. 
     Transmitters used in the wireless communication systems that are compliant with the aforementioned 802.11a/802.11g/802.11n standards as well as other standards such as the 802.16a/d/e/m IEEE Standards, typically perform multi-carrier OFDM symbol encoding (which may include error correction encoding and interleaving), convert the encoded symbols into the time domain using Inverse Fast Fourier Transform (IFFT) techniques, and perform digital to analog conversion and conventional radio frequency (RF) upconversion on the signals. These transmitters then transmit the modulated and upconverted signals after appropriate power amplification to one or more receivers, resulting in a relatively high-speed time domain signal with a large peak-to-average ratio (PAR). 
     Likewise, the receivers used in the wireless communication systems that are compliant with the aforementioned 802.11a/802.11g/802.11n and 802.16a IEEE standards typically include an RF receiving unit that performs RF downconversion and filtering of the received signals (which may be performed in one or more stages), and a baseband processor unit that processes the OFDM encoded symbols bearing the data of interest. The digital form of each OFDM symbol presented in the frequency domain is recovered after baseband downconverting, conventional analog to digital conversion and Fast Fourier Transformation of the received time domain analog signal. Thereafter, the baseband processor performs demodulation (phase rotation) and frequency domain equalization (FEQ) to recover the transmitted symbols, and these symbols are then processed in a Viterbi decoder to estimate or determine the most likely identity of the transmitted symbol. The recovered and recognized stream of symbols is then decoded, which may include deinterleaving and error correction using any of a number of known error correction techniques, to produce a set of recovered signals corresponding to the original signals transmitted by the transmitter. 
     In wireless communication systems, the RF modulated signals generated by the transmitter may reach a particular receiver via a number of different propagation paths, the characteristics of which typically change over time due to the phenomena of multi-path and fading. Moreover, the characteristics of a propagation channel differ or vary based on the frequency of propagation. To compensate for the time varying, frequency selective nature of the propagation effects, and generally to enhance effective encoding and modulation in a wireless communication system, each receiver of the wireless communication system may periodically develop or collect channel state information (CSI) for each of the frequency channels, such as the channels associated with each of the OFDM sub-bands discussed above. Generally speaking, CSI is information describing one or more characteristics of each of the OFDM channels (for example, the gain, the phase and the SNR of each channel). Upon determining the CSI for one or more channels, the receiver may send this CSI back to the transmitter, which may use the CSI for each channel to precondition the signals transmitted using that channel so as to compensate for the varying propagation effects of each of the channels. 
     An important part of a wireless communication system is therefore the selection of the appropriate data rates, and the coding and modulation schemes to be used for a data transmission based on channel conditions. Generally speaking, it is desirable to use the selection process to maximize throughput while meeting certain quality objectives, such as those defined by a desired frame error rate (FER), latency criteria, etc. 
     To further increase the number of signals that may be propagated in the communication system and/or to compensate for deleterious effects associated with the various propagation paths, and to thereby improve transmission performance, it is known to use multiple transmission and receive antennas within a wireless transmission system. Such a system is commonly referred to as a multiple-input, multiple-output (MIMO) wireless transmission system and is specifically provided for within the 802.11n IEEE Standard now being adopted. As is known, the use of MIMO technology produces significant increases in spectral efficiency and link reliability, and these benefits generally increase as the number of transmission and receive antennas within the MIMO system increases. 
     In addition to the frequency channels created by the use of OFDM, a MIMO channel formed by the various transmit and receive antennas between a particular transmitter and a particular receiver includes a number of independent spatial channels. As is known, a wireless MIMO communication system can provide improved performance (e.g., increased transmission capacity) by utilizing the additional dimensionalities created by these spatial channels for the transmission of additional data. Of course, the spatial channels of a wideband MIMO system may experience different channel conditions (e.g., different fading and multi-path effects) across the overall system bandwidth and may therefore achieve different SNRs at different frequencies (i.e., at the different OFDM frequency sub-bands) of the overall system bandwidth. Consequently, the number of information bits per modulation symbol (i.e., the data rate) that may be transmitted using the different frequency sub-bands of each spatial channel for a particular level of performance may differ from frequency sub-band to frequency sub-band. 
     However, instead of using the different transmit and receive antennas to form separate spatial channels on which additional information is sent, better reception properties can be obtained in a MIMO system by using each of the various transmit antennas of the MIMO system to transmit the same signal while phasing (and amplifying) this signal as it is provided to the various transmission antennas to achieve beamforming or beamsteering. Generally speaking, beamforming or beamsteering creates a spatial gain pattern having one or more high gain lobes or beams (as compared to the gain obtained by an omni-directional antenna) in one or more particular directions, while reducing the gain over that obtained by an omni-directional antenna in other directions. If the gain pattern is configured to produce a high gain lobe in the direction of each of the receiver antennas, the MIMO system can obtain better reception reliability between a particular transmitter and a particular receiver, over that obtained by single transmitter-antenna/receiver-antenna systems. 
     There are many known techniques for determining a steering matrix specifying the beamsteering coefficients that need to be used to properly condition the signals being applied to the various transmission antennas so as to produce the desired transmit gain pattern at the transmitter. As is known, these coefficients may specify the gain and phasing of the signals to be provided to the transmission antennas to produce high gain lobes in particular or predetermined directions. These techniques include, for example, transmit-MRC (maximum ratio combining) and singular value decomposition (SVD). An important part of determining the steering matrix is taking into account the specifics of the channel between the transmitter and the receiver, referred to herein as the forward channel. As a result, steering matrixes are typically determined based on the CSI of the forward channel. However, to determine the CSI or other specifics of the forward channel, the transmitter must first send a known test or calibration signal to the receiver, which then computes or determines the specifics of the forward channel (e.g., the CSI for the forward channel) and then sends the CSI or other indications of the forward channel back to the transmitter, thereby requiring signals to be sent both from the transmitter to the receiver and then from the receiver back to the transmitter in order to perform beamforming in the forward channel. Moreover, this exchange must occur each time the forward channel is determined (e.g., each time a steering matrix is to be calculated for the forward channel). This determination of the steering matrix based on the transmitting sending a calibration signal and receiving a CSI or other indication is known as explicit beamforming. 
     To reduce the amount of startup exchanges required to perform beamforming based on CSI or other channel information, it is known to perform implicit beamforming in a MIMO communication system. With implicit beamforming, the steering matrix is calculated or determined based on the assumption that the forward channel (i.e., the channel from the transmitter to the receiver in which beamforming is to be accomplished) can be estimated from the reverse channel (i.e., the channel from the receiver to the transmitter). In particular, the forward channel can ideally be estimated as the matrix transpose of the reverse channel. Thus, in the ideal case, the transmitter only needs to receive signals from the receiver to produce a steering matrix for the forward channel, as the transmitter can use the signals from the receiver to determine the reverse channel, and can simply estimate the forward channel as a matrix transpose of the reverse channel. As a result, implicit beamforming reduces the amount of startup exchange signals that need to be sent between a transmitter and a receiver because the transmitter can estimate the forward channel based solely on signals sent from the receiver to the transmitter. 
     Unfortunately, however, radio frequency (RF) chain impairments in the form of gain/phase imbalances and coupling losses impair the ideal reciprocity between the forward and the reverse channels, making it necessary to perform additional calibration exchanges each time the forward channel is being determined, to account for these impairments. In any event, these RF chain impairments render the use of implicit beamforming (which estimates the forward channel based solely on an estimate of the reverse channel) inferior in practice. 
     SUMMARY 
     In one embodiment, a method of beamforming within a communication system having a first transceiver device having a first plurality of antennas and a second transceiver device having a second plurality of antennas, the method comprises: identifying an index corresponding to one of the second plurality of antennas of the second transceiver; determining a partial dimensional description of a forward channel, wherein a signal travels from the first transceiver device to the second transceiver device via the forward channel; determining a cophase of the partial dimensional description of the forward channel at the index; determining a partial dimensional description of a reverse channel, wherein a signal travels from the second transceiver device to the first transceiver device via the reverse channel; determining a cophase of the partial dimensional description of the reverse channel at the index; developing a correction matrix from the cophase of the partial dimensional description of the reverse channel and the cophase of the partial dimensional description of the forward channel; using the correction matrix to process signals to be transmitted via the forward channel; and using a steering matrix to perform beamforming in the forward channel. 
     In another embodiment, a wireless transceiver for transmitting signals to one or more other communication devices, the wireless transceiver comprises: a multiplicity of antennas; a beamforming network coupled to the multiplicity of antennas; a controller coupled to the beamforming network to control the beamforming network using a steering matrix and to use a correction matrix to process signals to be transmitted via a forward channel; and a correction matrix calculation unit that obtains a reverse channel steering matrix and obtains a forward channel steering matrix, and develops the correction matrix from the reverse channel steering matrix and from the forward channel steering matrix; and a steering matrix calculation unit to develop (i) the steering matrix, (ii) the reverse channel steering matrix by taking a cophase of a partial dimensional description of the reverse channel, and (iii) the forward channel steering matrix by taking a cophase of a partial dimensional description of the forward channel. 
     In yet another embodiment, a method of beamforming within a communication system having a first transceiver with a first plurality of antennas and a second transceiver having a second plurality of antennas, the method comprising: transmitting a calibration initiation packet via a forward channel, the calibration initiation packet including a signal indicative of a request for a forward steering matrix feedback packet, wherein a signal travels from the first transceiver to the second transceiver via the forward channel; receiving via a reverse channel the forward steering matrix feedback packet, wherein a signal travels from the second transceiver to the first transceiver via the reverse channel; and determining a partial dimensional description of the reverse channel based on the reception of the forward steering matrix feedback packet; developing a reverse steering matrix feedback packet based on the partial dimensional description of the reverse channel; developing a correction matrix based the forward steering matrix feedback packet channel and the reverse steering matrix feedback packet; using the correction matrix to process signals to be transmitted via the forward channel; and using a steering matrix to perform beamforming in the forward channel. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of a wireless MIMO communication or transmission system that determines and uses a correction matrix as part of an implicit beamforming technique used in a transmitter of the MIMO communication system; 
         FIG. 2  is a flow diagram of an example method for calibrating a station in a wireless network; 
         FIG. 3  is a timing diagram illustrating communications between a Station A and a Station B during the calibration method of  FIG. 2 ; 
         FIG. 4  is a flow diagram of an example method for generating a correction matrix based on a partial estimation of a reverse channel and corresponding information from an estimation of a forward channel; 
         FIG. 5  is a flow diagram of another example method for generating a correction matrix based on determining forward and reverse steering matrices from a partial estimation of a reverse channel and corresponding information from an estimation of a forward channel; 
         FIG. 6  is a flow diagram of an example implementation of the method of  FIG. 5 ; 
         FIG. 7  is a block diagram of an example implementation of a space time mapping controller of  FIG. 1 ; 
         FIG. 8  is a flow diagram of another example method for calibrating a station in a wireless network; 
         FIG. 9  is a timing diagram illustrating communications between a Station A and a Station B during the calibration method of  FIG. 8 ; 
         FIG. 10  is a flow diagram of an alternative example method to that of  FIG. 8 ; 
         FIG. 11  is a flow diagram of another example method for calibrating a station in a wireless network; 
         FIG. 12  is a timing diagram illustrating communications between a Station A and a Station B during the calibration method of  FIG. 11 ; and 
         FIG. 13  is a block diagram illustrating a transmit gain pattern for wireless communications between a single transmitter and a single receiver implementing a transmitter beamforming technique that uses a calibration factor as part of an implicit beamforming technique. 
     
    
    
     DETAILED DESCRIPTION 
     While the beamforming techniques described herein for processing and effecting a wireless data transmission are described as being used in communication systems that use one of the IEEE Standard 802.11x communication standards, these techniques may be used in various other types of wireless communication systems and are not limited to those conforming to one or more of the IEEE Standard 802.11x standards. For example, these techniques may be used in communications based on the IEEE 802.16e, 802.16j, or 802.16m standards (known as “WiMAX”) and others. 
     Referring now to  FIG. 1 , a MIMO communication system  10  is illustrated in block diagram form as generally including a single transceiver device  12  (hereinafter referred to as transmitter  12 ) having multiple transmission antennas  14 A- 14 N and a single transceiver device  16  (hereinafter referred to as receiver  16 ) having multiple receiver antennas  18 A- 18 M. The number of transmission antennas  14 A- 14 N can be the same as, more than, or less than the number of receiver antennas  18 A- 18 M. As shown in  FIG. 1 , the transmitter  12  may include a controller  20  coupled to a memory  21 , a symbol encoder and modulator unit  22  and a space-time filtering or mapping block  24 , also referred to herein as a transmit beamforming network. The transmitter  12  may also include a matrix equalizer  25  and a symbol demodulator and decoder unit  26  to perform demodulation and decoding of signals received via the antennas  14 A- 14 N in a receive mode. Additionally, the transmitter  12  includes a steering matrix calculation unit  28  and a correction matrix calculation unit  29 . As will be understood, the processing applied at the transmitter  12  may be based on, for example, the CSI developed by the transmitter  12  in response to a reception of a test or control signal C R1  sent by the receiver  16 . In particular, a controller  40  or other unit within the receiver  16 , such as a channel determination unit  27 , may process the received control signal C R1  and develop therefrom a measured description of the reverse channel between the transmitter  12  and the receiver  16  by determining or characterizing the propagation effects of the reverse channel on the signal C R1  as it traveled through the reverse channel. 
     The controller  20  may be any desired type of controller and may be implemented as one or more standard multi-purpose, programmable processors, such as micro-processors, as application specific integrated circuits (ASICs), or may be implemented using any other desired types of hardware, software and/or firmware. The channel determination unit  27 , the steering matrix calculation unit  28  and the correction matrix calculation unit  29  may be implemented as one or more custom integrated circuits, ASICs, field programmable gate arrays (FPGAs), programmable logic arrays (PLAs), programmable processors, such as micro-processors or digital signal processing processors, or may be implemented using any other desired types of hardware, software and/or firmware. Likewise, the space-time mapping block  24  or beamforming network, and the matrix equalizer  25  may be implemented using known or standard hardware and/or software elements. If desired, various of the transmitter components, such as the controller  20 , the modulator unit  22 , the demodulator unit  26 , the channel determination unit  27 , the steering matrix calculation unit  28 , the correction matrix calculation unit  29 , the space-time mapping block  24  and the matrix equalizer  25  may be implemented in the same or in different hardware devices, such as in the same or different processors. Additionally, each of these components of the transmitter  12  may be disposed in a housing  31  (shown in dotted relief in  FIG. 1 ) and the routines or instructions for implementing the functionality of any of these components may be stored in the memory  21  or within other memory devices associated with the individual hardware used to implement these components. 
     During operation, information signals T x1 -T xn  which are to be transmitted from the transmitter  12  to the receiver  16  are provided to the symbol encoder and modulator unit  22  for encoding and modulation. Of course, any desired number of signals T x1 -T xn  may be provided to the modulator unit  22 , with this number generally being limited by the modulation scheme used by and the bandwidth associated with the MIMO communication system  10 . Additionally, the signals T x1 -T xn  may be any type of signals, including analog or digital signals, and may represent any desired type of data or information. Additionally, if desired, a known test or control signal C x1  (which may be stored in the memory  21 ) may be provided to the symbol encoder and modulator unit  22  for use in determining CSI related information describing the characteristics of the channel(s) between the transmitter  12  and the receiver  16 . If desired, the same control signal or a different control signal may be used to determine the CSI for each frequency and/or spatial channel used in the MIMO communication system  10 . 
     The symbol encoder and modulator unit  22  may interleave digital representations of the various signals T x1 -T xn  and C x1  and may perform any other known type(s) of error-correction encoding on the signals T x1 -T xn  and C x1  to produce one or more streams of symbols to be modulated and sent from the transmitter  12  to the receiver  16 . While the symbols may be modulated using any desired or suitable QAM technique, such as using 64 QAM, these symbols may be modulated in any other known or desired manner including, for example, using any other desired phase and/or frequency modulation techniques. In any event, the modulated symbol streams are provided by the symbol encoder and modulator unit  22  to the space-time mapping block  24  for processing before being transmitted via the antennas  14 A- 14 N. While not specifically shown in  FIG. 1 , the modulated symbol streams may be up-converted to the RF carrier frequencies associated with an OFDM technique (in one or more stages) before being processed by the space-time mapping block  24  in accordance with a beamforming technique more specifically described herein. Upon receiving the modulated signals, the space-time mapping block  24  or beamforming network processes the modulated signals by injecting delays and/or gains into the modulated signals based on a steering matrix provided by the controller  12 , to thereby perform beamsteering or beamforming via the transmission antennas  14 A- 14 N. 
     The signals transmitted by the transmitter  12  are received by the receiver antennas  18 A- 18 M and may be processed by a matrix equalizer  35  within the receiver  16  to enhance the reception capabilities of the antennas  18 A- 18 M. As will be understood, the processing applied at the receiver  16  (as well as at the transmitter  12 ) may be based on, for example, the CSI developed by the receiver  16  in response to the transmission of the test or control signal C x1 . In particular, a controller  40  or other unit within the receiver  16 , such as a channel determination unit  39 , may process the received control signal C x1  and develop therefrom a measured description of the forward channel between the transmitter  12  and the receiver  16  by determining or characterizing the propagation effects of the forward channel on the signal C x1  as the signal C x1  traveled through the forward channel. In any event, a symbol demodulator and decoder unit  36 , under control of the controller  40 , may decode and demodulate the received symbol strings as processed by the matrix equalizer  35 . In this process, these signals may be downconverted to baseband. Generally, the demodulator and decoder unit  36  may operate to remove effects of the forward channel based on the CSI as well as to perform demodulation on the received symbols to produce a digital bit stream. In some cases, if desired, the symbol demodulator and decoder unit  36  may perform error correction decoding and deinterleaving on the bit stream to produce the received signals R x1 -R xn  corresponding to the originally transmitted signals T x1 -T xn . 
     As shown in  FIG. 1 , the receiver  16  may also include a memory  41  and a symbol encoder and modulator unit  46  which may receive one or more signals T R1 -T Rm  which may be encoded and modulated using any desired encoding and modulation techniques. The receiver  16  may also provide one or more known test or control signals C R1  to the symbol encoder/modulator unit  46  to be sent to the transmitter  12  to enable the transmitter  12  to determine a measured description of the reverse channel between the receiver  16  and the transmitter  12 . The encoded and modulated symbol stream may then be upconverted and processed by a space-time mapping block  34  to perform beamsteering based on a steering matrix developed by a steering matrix calculation unit  48 , prior to being transmitted via the receiver antennas  18 A- 18 N to, for example, the transmitter  12 , thereby implementing the reverse link. As shown in  FIG. 1 , each of the receiver components may be disposed in a housing  51 . 
     The matrix equalizer  25  and the demodulator/decoder unit  26  within the transmitter  12  operate similarly to the matrix equalizer  35  and the demodulator/decoder unit  36  of the receiver  16  to demodulate and decode the signals transmitted by the receiver  16  to produce the recovered signals R R1 -R Rm . Here again, the matrix equalizer  25  may process the received signals in any known manner to enhance the separation and therefore the reception of the various signals transmitted by the antennas  18 A- 18 M. Of course, the CSI or other measured description of the forward channel for the various OFDM channel(s) may be used by the steering matrix calculation units  28  and  48  as well as by the controllers  20  and  40  to perform beamforming and to determine a steering matrix used by the space-time mapping blocks  24 ,  34 . As noted above, the CSI, beamforming and other programs and data such as the steering matrix used by the units  28  and  48  and by the controllers  20  and  40 , a correction matrix determined by the correction matrix calculation unit  29 , etc. may be stored in the memories  21  and  41 . 
     As is generally known, beamforming or beamsteering typically includes applying appropriate phases and gains to the various signals as sent through the multiple transmission antennas  14 A- 14 N, in a manner which causes the signals sent from the different transmission antennas  14 A- 14 N to constructively interact (add in phase) in certain predetermined directions and to deconstructively interact (cancel) in other directions. Thus, beamsteering typically produces a beam pattern having high gain regions (referred to as high gain lobes) in various predetermined directions and low gain regions (typically referred to as nulls) in other directions. The use of beamforming techniques in a MIMO system enables a signal to be sent with high gain (as compared to an omni-directional antenna) in certain directions, and to be sent with low gain (as compared to an omni-directional antenna) in other directions. Thus, in the MIMO system  10  of  FIG. 1 , beamforming may be used to enhance signal directivity towards the receiver antennas  18 A- 18 M, which improves the SNR of the transmissions and results in more reliable transmissions. In this case, the beamforming technique will generally form high gain lobes in the direction of propagation at which the highest gain is desired, and in particular in the directions of propagation from the transmitter  12  to each of the receiver antennas  18 A- 18 M of the receiver  16  or to the receiver  16  in general. 
     To implement beamforming in the transmitter  12 , the steering matrix calculation unit  28  may determine or calculate a set of matrix coefficients (referred to herein as a steering matrix) which are used by the space-time mapping block or beamforming network  24  to condition the signals being transmitted by the antennas  14 A- 14 N. Generally speaking, the steering matrix for any particular frequency channel of the MIMO system  10  (in the forward channel between the transmitter  12  and the receiver  16 ) may be determined by the steering matrix calculation unit  28  based on the CSI determined for that forward channel. In this case, the steering matrix calculation unit  28  may use any desired beam steering or matrix computation techniques, such as transmit-MRC or SVD techniques, to compute the steering matrix. As these techniques are well known in the art, they will not be discussed in detail herein. 
     However, as is known, to actually determine the CSI or other measured description of the forward channel, i.e., for the channel from the transmitter  12  to the receiver  16 , the transmitter  12  generally sends a known control or test signal to the receiver  16  (e.g., the signal C x1 ) and the receiver  16  may then determine the CSI or other measured description of the forward channel and send this information back to the transmitter  12  as part of a payload of a transmission. In the event of explicit beamforming, in this case, the transmitter  12  must first send a test or control signal to the receiver  16 , which then determines a measured description of the forward channel and sends this description of the forward channel from the receiver  16  back to the transmitter  12 . This characterization of the forward channel thereby requires, each time the steering matrix is computed, multiple communications between the transmitter  12  and the receiver  16  so as to enable the transmitter  12  to obtain the CSI or other description of the forward channel used to develop the steering matrix to be used in the forward channel. In explicit transmit beamforming, RF chain imbalance should not impact the beamforming performance, as the forward channel is explicitly known. Additionally, in the case of implicit beamforming, to avoid the use of multiple communication between a particular transmitter/receiver pair each time a steering matrix is to be computed for the forward channel, the transmitter  12  may determine the CSI or other measured description of the reverse channel, i.e., the channel from the receiver  16  to the transmitter  12 , from the signal(s) sent from the receiver  16  including, for example the known test or control signal C R1 . Based on the CSI or other measured description of the reverse channel, the transmitter  12  may calculate the steering matrix for the forward channel. 
     To reduce or account for the errors introduced by RF chain impairments in a standard implicit beamforming technique, the transmitter  12  may use a calibration technique that applies a correction matrix during the beamforming process to compensate for measured differences between the actual forward and reverse channels. In particular, this technique first determines a correction matrix as a function of measured descriptions of the forward and the reverse channels. Then, each time a new steering matrix is to be calculated for the forward channel, the beamforming technique applies the correction matrix to a steering matrix determined using a basic implicit beamforming technique, so that, once the correction matrix is determined, the transmitter may simply perform implicit beamforming using a measured description of the reverse channel (i.e., the channel between the receiver and the transmitter) to produce an estimate of the forward channel (i.e., the channel between the transmitter and the receiver). Alternatively, the transmitter  12  may also calculate correction matrices for its receive chains, so that once the correction matrix is determined, the transmitter may apply it to the reverse channel (i.e., the channel from the receiver  16  to the transmitter  12 ) estimation, and perform implicit beamforming using a measured description of this processed reverse channel estimate to produce an estimate of the forward channel (i.e., the channel from the transmitter  12  to the receiver  16 ). The calibration procedure may be conducted infrequently, compared with steering matrix updates. For example, it may be conducted only upon association of the device into the network, or upon the changes in the environment (e.g. a change in temperature). 
     Transmission from the transmitter  12  (Station A) to the receiver  16  (Station B) can be modeled as:
 
 y   B   ={tilde over (H)}   AB   Q   A   X   A   +n   B ,  (Equ. 1)
 
where y B  and n B  are the received signal vector and additive noise vector at Station B, respectively; {tilde over (H)} AB  is the equivalent channel from Station A to Station B; x A  is the signal vector to be transmitted from Station A; and Q A  is the steering matrix (which may be a vector) at Station A that spreads the signal vector onto actual transmitting chains at Station A. Q A  may be designed based on the knowledge of {tilde over (H)} AB  at station A using. In the transmitter  12 , the steering matrix Q A  may be determined by the steering matrix calculation unit  28  based, for example, on the CSI determined for the channel from the transmitter  12  to the receiver  16 . The steering matrix Q A  may be determined using a variety of techniques, including techniques known to those of ordinary skill in the art
 
     In implicit transmit beamforming, Station A determines an estimation of {tilde over (H)} AB  based on an estimate of the channel from Station B to Station A. In the case of time-division duplexing (TDD), the forward link and the reverse link share the same frequency band, so their physical propagation channels, denoted as H AB  and H BA  respectively, can be assumed reciprocal (H AB =H BA   T ), if the channel is varying slowly compared to the interval between forward and reverse link transmissions. 
     However, the actual channels observed at baseband also include the equivalent radio frequency (RF) responses of the transmit and receive chains, which might not be identical for the transmit and receive chains in the same device. This imbalance results in the actual channels {tilde over (H)} AB  and {tilde over (H)} BA  not being reciprocal. The mathematical description of this imbalance issue may be represented as:
 
 {tilde over (H)}   AB   =C   B,Rx   H   AB   C   A,Tx ,  (Equ. 2)
         where C B,Rx  represents the RF responses at the receive chains of Station B; and where C A,Tx  represents the RF responses at the transmit chains of Station A. By ignoring the coupling among transmit and receive chains, the matrices C B,Rx  and C B,Rx  can be approximately modeled as diagonal matrices.       

     The equivalent channel from Station B to Station A, {tilde over (H)} BA , can be represented as:
 
 {tilde over (H)}   BA   =C   A,Rx   H   AB   T   C   B,Tx ,  (Equ. 3)
         Due to imbalances between the transmit and receive chains at Station A and at Station B, respectively, i.e., C A,Tx ≠C A,Rx   T  and/or C B,Tx ≠C B,Rx   T , it stands that {tilde over (H)} AB ≠{tilde over (H)} BA   T .       

     One or both devices may compensate the transmit and the receiver RF imbalance at baseband. For example, one or more correction matrices may be calculated and then multiplied with the transmitted or received signal vector in the baseband to correct the imbalance and maintain reciprocity between the transmit and receive channels. For example, a transmitter-side correction matrix K A,Tx  and a receiver-side correction matrix K B,Rx  are calculated such that they completely compensate for the imbalance. Thus, a corrected equivalent channel from Station A to Station B Ĥ AB  may be represented as:
 
 Ĥ   AB   =K   B,Rx   {tilde over (H)}   AB   K   A,Tx   =α{tilde over (H)}   BA   T ,  (Equ. 4)
         where α can be any scalar. Equation 4 is indicative of a system that may be referred to as a strict reciprocity system, i.e., a system that offers full calibration at both the transmitter and receiver. To implement a strict reciprocity system, Station A left-multiplies the correction matrix K A,Tx  with the signal it is to transmit (Q A x A ) at baseband. Also, upon receiving the signal, Station B left-multiplies the correction matrix K B,Rx  with the received signal at baseband.       

     An alternative approach is to calculate the correction matrices for the reverse channel K A,Rx  and/or K B,Tx , such that:
 
 Ĥ   BA   =K   A,Rx   {tilde over (H)}   BA   K   B,Tx   =α{tilde over (H)}   AB   T ,  (Equ. 5)
         where α′ can be any scalar. To implement strict reciprocity, Station B left-multiplies the correction matrix K B,Tx  with the reverse channel sounding signal Station B is to transmit at baseband. Upon receiving the signal, Station A left-multiplies the correction matrix K A,Rx  with the estimated reverse channel at baseband.       

     Many implicit transmit beamforming methods only utilize transmitter-side compensation. In these cases, Ĥ AB  may be represented as:
 
 Ĥ   AB   ={tilde over (H)}   AB   K   A,Tx   =D   B   {tilde over (H)}   BA   T ,  (Equ. 6)
         where D B  is a diagonal matrix representing the imbalance at Station B. Equation 6 is indicative of a system that may be referred to as a semi-reciprocal system, in this example, because it offers partial calibration on the transmitter. With semi-reciprocity, only transmitter-side compensation need be utilized. To implement a semi-reciprocal system, Station A left-multiplies the correction matrix K A,Tx  with the signal it is to transmit (Q A x A ) at baseband. Upon receiving the signal, Station B need not apply a correction matrix, such as the correction matrix K B Rx , with the received signal. Similarly, the transmitter-side compensation may also be applied at the receiver chains of the same device. That is, to realize semi-reciprocity, Station A left-multiplies the correction matrix K A,Rx  with the reverse channel estimates at baseband:
 
 Ĥ   BA   =K   A,Rx   {tilde over (H)}   BA   (Equ. 7)
 
so,
 
 Ĥ   AB   =D′   B   {tilde over (H)}   BA   T   (Equ. 8)
   where K A,Rx =inv(K A,Tx ). Here we use the forward channel correction (c.f. Equations 1, 2, 3, 4, 5) to illustrate the proposed calibration methods. The extension to reverse channel correction matrix will be straightforward to one of ordinary skill in the art.       

       FIG. 2  is a flow diagram of an example method  200  for calibrating a station in a wireless network using an explicit beamforming technique. The method  200  will be described with reference to  FIG. 3 , which is a timing diagram illustrating communications between a Station A and a Station B during the calibration method  200 . At a block  204 , Station A initiates a calibration process by sending a packet  208  to Station B. The packet  208  sent at the block  204  includes a “calibration initiation” signal, which indicates to Station B that Station A is requesting to initialize calibration. The calibration initiation signal may be sent at a physical layer or a media access control layer, for example, of a wireless communication protocol. The packet  208  may be a sounding packet, for example. 
     At a block  212 , in response to receiving the packet  208  with the calibration initiation signal, Station B generates a full-dimensional channel state information (CSI) estimation of {tilde over (H)} AB . At a block  214 , after Station B generates the full-dimensional CSI estimation of {tilde over (H)} AB  (block  212 ), Station B transmits it back to Station A via a CSI feedback packet  216 . According to many wireless network protocols, transmission of the CSI feedback packet  216  typically is not considered time critical. 
     To generate the CSI estimation, Station B sets its spatial steering matrix Q B  to a predetermined N ss     —     B ×N ss     —     B  square matrix, such as the identity matrix I, a Hadamard matrix, a discrete Fourier transform (DFT) matrix, etc., where N ss     —     B  is the number of data streams transmitted by the CSI feedback packet  216 , which could be less than the number of transmit antennas at Station B. For example, using the identity matrix I, Station B uses only a portion of its available antennas to send the CSI feedback packet  216 , e.g., using the same number of antennas as there are to be spatial streams in the CSI feedback packet  216 . In some examples, Station B will use the same data rate and bandwidth setting (e.g., as set forth in the 802.11n standard) as are in the calibration initial packet (e.g., sounding packet)  208 , when sending the CSI feedback packet  216 . The CSI feedback packet  216  is a not a sounding packet because it does not include training information for all available spatial dimensions of a multiple-antenna channel. Rather, the CSI feedback packet  216  includes one or more characteristics of {tilde over (H)} AB , e.g., of each of the OFDM channels (for example, the gain, the phase and the SNR of each channel). Such a packet may be referred to as a non-sounding packet, although in the case where Station B uses a single receiver side antenna to transmit the reverse channel CSI feedback packet  216 , that packet essentially provides a full sounding of the antenna used. 
     At a block  218 , after receiving the CSI feedback packet  216 , Station A calculates the correction matrix K A,Tx . For example, at a block  220 , from the received CSI feedback packet  216 , Station A determines a partial estimate of the reverse channel [{tilde over (H)} BA ] *Ψ     B   , where Ψ B  indicates the subset of the antennas at Station B. In response to this partial estimation (block  220 ) and the full-dimension CSI estimation (block  214 ), the block  218  calculates the correction matrix K A,Tx . In other examples, Station A additionally or alternatively calculates a correction matrix K A,Rx  e.g., at the block  218 . Example methods for generating the correction matrices K A,Tx  and K A,Rx  will be described below. As shown, even if Station B uses only a subset of its antennas in sending the CSI feedback packet  216 , Station A is able to determine the correction matrix, K A,Tx . 
     In some examples, the method  200  of  FIG. 2  is implemented by the system  10  of  FIG. 1 . Of course, in other examples, the method  200  is implemented by other systems. Also, the system  10  need not implement the method  200 , but may alternatively implement other methods, includes other methods described herein. 
     In an example, the controller  20  causes the packet  208  to be sent to the receiver  16  (block  204 ). The controller  40  causes the CSI feedback packet  216  to be sent to the transmitter  12  (block  214 ). The channel determination unit  39  generates the full-dimensional CSI estimation of {tilde over (H)} AB  (block  212 ), and the controller  40  causes this information to be sent back to the transmitter (block  214 ). The correction matrix calculation unit  29  generates [{tilde over (H)} BA ] *Ψ     B    (block  220 ). The correction matrix calculation unit  29  also generates the one or more correction matrices (block  218 ). The space-time mapping block  24  left-multiplies the correction matrix K A,Tx  with the signal to be transmitted at baseband. Optionally, the steering matrix calculation block  28  modifies the steering matrix Q A  using the correction matrix K A,Tx  by, for example, left-multiplying the correction matrix K A,Tx  with the steering matrix Q A  to generate a modified steering matrix. Alternatively, the correction matrix calculation unit  29  modifies the steering matrix Q A  using the correction matrix K A,Tx . In other examples, the matrix equalizer  25  left-multiplies the correction matrix K A,Rx  with the signal it receives from the station  16  at baseband. 
       FIG. 4  is a flow diagram of an example method  300  for generating a correction matrix, K A,Tx  or K A,Rx , based on the partial estimation of the reverse channel {tilde over (H)} BA  (i.e., [{tilde over (H)} BA ] *Ψ     B   ) and the partial estimation of the forward channel {tilde over (H)} AB  (i.e., [{tilde over (H)} AB ] *Ψ     B   ). In this example method, it is assumed that the correction matrix to be generated, K A,Tx , is a diagonal matrix. 
     At a block  304 , for iεΨ B , a corresponding estimate of K A,Tx  may be calculated based on the i-th row in [{tilde over (H)} AB ] *Ψ     B    and the i-th row in [{tilde over (H)} BA   T ] *Ψ     B   . The k-th element of the i-th row of [{tilde over (H)} AB ] *Ψ     B    may be written as:
 
 [{tilde over (H)}   AB ] ki   =c   B,Rx     —     i   [H   AB ] ik   C   A,Tx     —     k ,  (Equ. 9)
         Similarly, the k-th element of the i-th row of [{tilde over (H)} BA   T ] *Ψ     B    is the k-th element of the i-th column of [{tilde over (H)} BA ] *Ψ     B   , which may be written as:
 
 [{tilde over (H)}   BA ] ki   =c   A,Rx     —     k   [H   AB ] ik   c   B,Tx     —     i ,  (Equ. 10)
   The (k,k) element of the diagonal correction matrix K A,Tx  corresponding to i may be determined as:       

     
       
         
           
             
               
                 
                   
                     
                       
                         [ 
                         
                           K 
                           
                             A 
                             , 
                             Tx 
                           
                         
                         ] 
                       
                       kk 
                     
                     = 
                     
                       
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               BA 
                             
                             ] 
                           
                           ki 
                         
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               AB 
                             
                             ] 
                           
                           ik 
                         
                       
                       = 
                       
                         
                           
                             ( 
                             
                               
                                 c 
                                 
                                   B 
                                   , 
                                   Tx_i 
                                 
                               
                               
                                 c 
                                 
                                   B 
                                   , 
                                   Rx_i 
                                 
                               
                             
                             ) 
                           
                           ⁢ 
                           
                             
                               c 
                               
                                 A 
                                 , 
                                 Rx_k 
                               
                             
                             
                               c 
                               
                                 A 
                                 , 
                                 Tx_k 
                               
                             
                           
                         
                         = 
                         
                           α 
                           ⁢ 
                           
                             
                               c 
                               
                                 A 
                                 , 
                                 Rx_k 
                               
                             
                             
                               c 
                               
                                 A 
                                 , 
                                 Tx_k 
                               
                             
                           
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     11 
                   
                   ) 
                 
               
             
           
         
       
         
         
           
             where 
           
         
       
    
     
       
         
           
             
               
                 
                   
                     α 
                     ⁢ 
                     
                       
                         c 
                         
                           B 
                           , 
                           
                             T 
                             ⁢ 
                             x_i 
                           
                         
                       
                       
                         c 
                         
                           B 
                           , 
                           Rx_i 
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     12 
                   
                   ) 
                 
               
             
           
         
       
         
         
           
             Thus, for example, the (1,1) element of K A,Tx  corresponding to i may be determined as: 
           
         
       
    
     
       
         
           
             
               
                 
                   
                     
                       
                         [ 
                         
                           K 
                           
                             A 
                             , 
                             Tx 
                           
                         
                         ] 
                       
                       11 
                     
                     = 
                     
                       
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               BA 
                             
                             ] 
                           
                           
                             1 
                             ⁢ 
                             i 
                           
                         
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               AB 
                             
                             ] 
                           
                           
                             i 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             1 
                           
                         
                       
                       = 
                       
                         
                           
                             ( 
                             
                               
                                 c 
                                 
                                   B 
                                   , 
                                   Tx_i 
                                 
                               
                               
                                 c 
                                 
                                   B 
                                   , 
                                   Rx_i 
                                 
                               
                             
                             ) 
                           
                           ⁢ 
                           
                             
                               c 
                               
                                 A 
                                 , 
                                 
                                   Rx_ 
                                   ⁢ 
                                   1 
                                 
                               
                             
                             
                               c 
                               
                                 A 
                                 , 
                                 
                                   Tx_ 
                                   ⁢ 
                                   1 
                                 
                               
                             
                           
                         
                         = 
                         
                           α 
                           ⁢ 
                           
                             
                               c 
                               
                                 A 
                                 , 
                                 
                                   Rx_ 
                                   ⁢ 
                                   1 
                                 
                               
                             
                             
                               c 
                               
                                 A 
                                 , 
                                 
                                   Tx_ 
                                   ⁢ 
                                   1 
                                 
                               
                             
                           
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     13 
                   
                   ) 
                 
               
             
           
         
       
         
         
           
             Similarly, the (2,2) element of K A,Tx  corresponding to i may be determined as: 
           
         
       
    
     
       
         
           
             
               
                 
                   
                     
                       
                         [ 
                         
                           K 
                           
                             A 
                             , 
                             Tx 
                           
                         
                         ] 
                       
                       22 
                     
                     = 
                     
                       
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               BA 
                             
                             ] 
                           
                           
                             2 
                             ⁢ 
                             i 
                           
                         
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               AB 
                             
                             ] 
                           
                           
                             i 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             2 
                           
                         
                       
                       = 
                       
                         α 
                         ⁢ 
                         
                           
                             c 
                             
                               A 
                               , 
                               
                                 Rx_ 
                                 ⁢ 
                                 2 
                               
                             
                           
                           
                             c 
                             
                               A 
                               , 
                               
                                 Tx_ 
                                 ⁢ 
                                 2 
                               
                             
                           
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     14 
                   
                   ) 
                 
               
             
           
         
       
         
         
           
             As can be seen from Equations 11-14, Ĥ AB ={tilde over (H)} AB K A,Tx =αC B,Rx =αD B {tilde over (H)} BA   T , where D B =C B,Rx C B,Tx   −1 , and the calibrated channels are semi-reciprocal. 
           
         
       
    
     At a block  308 , the correction matrix K A,Tx  is generated based on the i estimates generated at the block  304 . As just one example, the correction matrix K A,Tx  could be generated base on an average of the i estimates. For instance, the (1,1) element is generated as an average of the i estimates of the (1,1) element generated at the block  304 . Similarly, the (2,2) element is generated as an average of the i estimates of the (2,2) element generated at the block  304 , etc. 
     The correction matrix K A,Rx  is calculated by inverting the correction matrix K A,Tx . Alternatively, the correction matrix K A,Rx  is calculated according to a method similar to the method  300  and equations similar to Equations 8 and 9. Then, the correction matrix K A,Tx  is calculated by inverting the correction matrix K A,Rx . 
     In some examples, block  308  determines only the phase mismatch in calculating the correction matrix K A,Tx  for example by averaging over different antenna index i or over different OFDM sub-carriers. In such examples, the diagonal correction matrix K A,Tx  is expressed as a phase correction factor where the phase difference between the reverse estimation channel and the forward estimation channel is determined to provide the phase difference for each antenna k of Station A:
 
[ K   A,Tx ] kk =exp{ j∠[{tilde over (H)}   BA ] ki   −∠[{tilde over (H)}   AB ] ik )},∀ i   (Equ. 15)
 
       FIG. 5  is a flow diagram of another example method  400  for generating a correction matrix based on the partial estimation of the reverse channel {tilde over (H)} BA  (i.e., [{tilde over (H)} BA ] *Ψ     B   ) and the partial estimation of the forward channel {tilde over (H)} AB  (i.e., [{tilde over (H)} AB ] *Ψ     B   ). 
     Block  402  determines the singular matrixes that define the estimate of the forward channel [{tilde over (H)} AB ] *Ψ     B    and the estimate of the reverse channel [{tilde over (H)} BA ] *Ψ     B   . A singular value decomposition (SVD) method or any other method or technique that determines a set of right singular matrixes that accurately describes or defines the forward channel [{tilde over (H)} AB ] *Ψ     B    and another set of right singular matrixes that accurately describes or defines the estimate of the reverse channel [{tilde over (H)} BA ] *Ψ     B    may be used. Or, in other examples, the estimate of the reverse channel [{tilde over (H)} BA   T ] *Ψ     B    is determined from inverting the estimate of the forward channel [{tilde over (H)} AB ] *Ψ     B   . A block  404  then determines a forward steering matrix, V F , using a method f, and a reverse steering matrix, V R , using the same method f and computes the correction matrix using:
 
 V   F   =f ([ {tilde over (H)}   AB ] *Ψ     B   ),  (Equ. 16)
 
 V   R   =f ([ {tilde over (H)}   BA   T ] *Ψ     B   )  (Equ. 17)
         Block  406  then determines the correction matrix based on the determined forward and backward steering matrices, according to:
 
 K   A,Tx   =V   F   V   R   H ,  (Equ. 18)
       

     In some examples, the diagonal of the calculated correction matrix is made to correspond to pure phase shifts by normalization. For example, the block  406  may express the correction matrix with phase correction only by applying the mathematical formula:
 
 K   A,Tx =exp{ j ∠( V   F   V   R   H )}  (Equ. 19)
 
     While the foregoing describes determining the correction matrix K A,Tx , the correction matrix K A,Rx  can be calculated instead, e.g., by inverting the correction matrix K A,Tx  determined using Equations 14, 15, 18, or 19. Alternatively, the correction matrix K A,Rx  is calculated first according to equations similar Equations 12-19. Then, the correction matrix K A,Tx  is calculated by inverting the correction matrix K A,Rx . 
       FIG. 6  illustrates an example approach  500  to the method  400 , and as may be achieved by the blocks  404  and  406  in  FIG. 5 . Specifically, the block  404  determines the forward steering matrix, V F , and the reverse steering matrix, V R , using a cophasing technique. A block  502  selects one Station B antenna index i. For example, Station A selects the index i corresponding to the Station B antenna with the strongest indicated channel gain from the CSI feedback packer  216 . In another example, Station A selects the antenna index i based on row averages. 
     With the generated partial estimates of the forward and reverse channels, block  504  determines the forward steering matrix V F  by taking a cophasing of the partial estimation [{tilde over (H)} AB ] *i , as indicated by: 
     
       
         
           
             
               
                 
                   
                     v 
                     F 
                   
                   = 
                   
                     
                       cophase 
                       ⁡ 
                       
                         ( 
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               AB 
                             
                             ] 
                           
                           
                             i 
                             * 
                           
                         
                         ) 
                       
                     
                     = 
                     
                       
                         cophase 
                         ⁡ 
                         
                           ( 
                           
                             [ 
                             
                               
                                 h 
                                 
                                   i 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   1 
                                 
                                 
                                   ( 
                                   F 
                                   ) 
                                 
                               
                               ⁢ 
                               
                                 h 
                                 
                                   i 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   2 
                                 
                                 
                                   ( 
                                   F 
                                   ) 
                                 
                               
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               … 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 h 
                                 
                                   iN 
                                   Tx 
                                 
                                 
                                   ( 
                                   F 
                                   ) 
                                 
                               
                             
                             ] 
                           
                           ) 
                         
                       
                       = 
                       
                         
                           [ 
                           
                             
                               
                                 
                                   ⅇ 
                                   
                                     jθ 
                                     
                                       NTx 
                                       1 
                                     
                                   
                                 
                               
                             
                             
                               
                                 
                                   ⅇ 
                                   
                                     jθ 
                                     
                                       NTx 
                                       2 
                                     
                                   
                                 
                               
                             
                             
                               
                                 ⋮ 
                               
                             
                             
                               
                                 1 
                               
                             
                           
                           ] 
                         
                         . 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     20 
                   
                   ) 
                 
               
             
           
         
       
     
     The block  504  also determines the reverse steering matrix V R  by taking a cophasing of the partial estimation [{tilde over (H)} BA   T ] *i , as indicated by: 
     
       
         
           
             
               
                 
                   
                     v 
                     R 
                   
                   = 
                   
                     
                       cophase 
                       ⁡ 
                       
                         ( 
                         
                           
                             [ 
                             
                               
                                 H 
                                 ~ 
                               
                               BA 
                               T 
                             
                             ] 
                           
                           
                             i 
                             * 
                           
                         
                         ) 
                       
                     
                     = 
                     
                       
                         cophase 
                         ⁡ 
                         
                           ( 
                           
                             [ 
                             
                               
                                 h 
                                 
                                   i 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   1 
                                 
                                 
                                   ( 
                                   R 
                                   ) 
                                 
                               
                               ⁢ 
                               
                                 h 
                                 
                                   i 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   2 
                                 
                                 
                                   ( 
                                   R 
                                   ) 
                                 
                               
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               … 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 h 
                                 
                                   iN 
                                   Tx 
                                 
                                 
                                   ( 
                                   R 
                                   ) 
                                 
                               
                             
                             ] 
                           
                           ) 
                         
                       
                       = 
                       
                         
                           [ 
                           
                             
                               
                                 
                                   ⅇ 
                                   
                                     jΨ 
                                     
                                       NTx 
                                       1 
                                     
                                   
                                 
                               
                             
                             
                               
                                 
                                   ⅇ 
                                   
                                     jΨ 
                                     
                                       NTx 
                                       2 
                                     
                                   
                                 
                               
                             
                             
                               
                                 ⋮ 
                               
                             
                             
                               
                                 1 
                               
                             
                           
                           ] 
                         
                         . 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     Equ 
                     . 
                     
                         
                     
                     ⁢ 
                     21 
                   
                   ) 
                 
               
             
           
         
       
     
     By block  502  identifying only a single receive antenna then the channel reduces to a single row, leaving each of V F  and V R  as linear matrices. Block  506 , applying Equation 18, produces the correction matrix K A,Tx , representing the phase difference between each of the corresponding forward and reverse steering matrices: 
     
       
         
           
             
               K 
               
                 A 
                 , 
                 Tx 
               
             
             = 
             
               
                 
                   V 
                   F 
                 
                 ⁢ 
                 
                   V 
                   R 
                   H 
                 
               
               = 
               
                 [ 
                 
                   
                     
                       
                         ⅇ 
                         
                           j 
                           ⁡ 
                           
                             ( 
                             
                               
                                 θ 
                                 
                                   NTx 
                                   1 
                                 
                               
                               - 
                               
                                 Ψ 
                                 
                                   NTx 
                                   1 
                                 
                               
                             
                             ) 
                           
                         
                       
                     
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                   
                   
                     
                       
                           
                       
                     
                     
                       
                         ⅇ 
                         
                           j 
                           ⁡ 
                           
                             ( 
                             
                               
                                 θ 
                                 
                                   NTx 
                                   2 
                                 
                               
                               - 
                               
                                 Ψ 
                                 
                                   NTx 
                                   2 
                                 
                               
                             
                             ) 
                           
                         
                       
                     
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                   
                   
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                     
                       ⋱ 
                     
                     
                       
                           
                       
                     
                   
                   
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                     
                       
                           
                       
                     
                     
                       1 
                     
                   
                 
                 ] 
               
             
           
         
       
     
     In terms of calibration architecture, at an implicit calibration beamformer, any of the determinations made above can be performed by the corresponding steering matrix calculator  28  or calibration factor calculator  29 , respectively, and performed in either software or hardware or a combination thereof. That is, the forward CSI matrix sent by Station B and the reverse SCI matrix determined by Station A may be sent to either software or hardware within Station A to determine the correction matrices. When steering a signal using the implicit calibration beamformer computed steering matrix, the implicit beamformer may apply the transmitter side calibration matrix K A,Tx  on the transmitted steered packets when multiplying the computer steering matrix with the signal stream, using the following expression (which is similar to Equ. 1):
 
 y   B   =H   AB   K   A,Tx   V   steer   x   A   +n   B ,  (Equ. 22)
         That is, Station A left-multiplies K A,Tx  with the signal it is to transmit (Q A x A ) at baseband, where Q A  is the steering matrix, V steer , (which may be a vector) at Station A that spreads the signal vector onto actual transmitting chains at Station A at baseband. Optionally, Station A modifies the steering matrix Q A  with the correction matrix by, for example, left-multiplying K A,Tx  with Q A .       

     Alternatively, Station A applies a receiver side calibration matrix K A,Rx  on the reverse channel estimation, i.e., from CSI feedback packet  216 , based on which the steering matrix is computed according to the following:
 
 {tilde over (H)}   BA   =K   A,Rx   {tilde over (H)}   BA ,  (Equ. 7)
 
 V   steer   =f ( Ĥ   BA   T ),  (Equ. 23)
         Here, Station A left-multiplies K A,Rx  with the signal it receives from Station B (y A ={tilde over (H)} BA Q B x B +n A , where y A  and n A  are the received signal vector and additive noise vector at Station A, respectively; {tilde over (H)} BA  is the equivalent channel from Station B to Station A; x B  is the signal vector to be transmitted from Station B; and Q B  is the steering matrix (which may be a vector) at Station B that spreads the signal vector onto actual transmitting chains at Station B at baseband).       

     In either case, multiplication of the correction matrix, whether performed in software or hardware, is performed in the frequency domain, for example, multiplying at each OFDM sub-carrier in the frequency domain, in the time domain, or in a combination of both. 
       FIG. 7  illustrates an example configuration  600  of the space-time mapping blocks  24  and/or  34  of  FIG. 1 , which apply the determined correction matrix to the forward and/or reverse channels accordingly. The spatial streams, x, are provided to a beamsteering matrix controller  602 , which applies the computed steering matrix, V steer , computed as discussed above and stored in a memory  604 . When compensating only the phase differences among the antennas on the Station A, the controller  602  provides parallel output signals to a phase correction stage  606 , which then applies different phase shifts for each output signal, as determined by the stored correction matrix (K A,Tx )  609 , e.g., by only using the phase components unless the correction matrix is only expressed in terms of phase. The stage  606  may be implemented by different phase shifter stages  602   a - 602   d  (not individually labeled), one for each transmitter chain (antenna). 
     In the illustrated example, correction of the phase shift is determined in the frequency domain. The correction occurs at the transmitter of Station A, as shown, or in other examples, at the receiver of Station B. At the transmitter side, the correction is performed on each sub-carrier at each transmitter chain, corresponding to each antenna, after the beamsteering matrix, V steer , has been applied to the signal stream, x. 
     In some examples, Station A also compensates for gain mismatches, by using a gain correction stage  608  implemented, for example, using a digital variable gain amplifier (DVGA). The stage is implemented by different DVGA gain stages  608   a - 608   d  (not individually labeled), one for each transmit antenna, in the illustrated example. The gain correction stage applies digital gain with correction values, from the correction matrix, K A,Tx , to each transmitter chain, where the corresponding matrix multiplication may be done in the time domain. In the illustrated example, an inverse fast Fourier transform (IFFT) controller  610  converts the phase corrected signal of stage  606  from the frequency domain to the time domain for the gain correction stage  608 . A corresponding DAC converter stage  612  provides the corrected signal stream to the corresponding antennas of Station A. 
     In some configurations, a MIMO-OFDM transmitter applies time domain cyclic delay diversity (TCDD) after the IFFT and CP insertion of block  610  on the transmitter chains. In some calibration procedures, the Station A turns OFF the TCDD function whenever sending a calibration initiation (e.g. sounding) packet and whenever Station A transmits the steered data packets determined using an implicit beamforming technique. If Station A does not turn OFF TCDD, then in other examples Station A manually applies the TCDD effects (i.e., linear phase shifts across sub-carriers) in the frequency domain of the channel estimations on the packets of the reverse channel. 
       FIG. 8  illustrates another method  700  for calibration of communications between Station A and Station B. The method  700  is described in reference to the timing diagram illustrated in  FIG. 9 . At a block  704 , Station A initiates a calibration process by sending a packet  708  to Station B. The packet  708 , like the packet  208 , may include a “calibration initiation” signal, which indicates to Station B that Station A is requesting to initialize calibration. For the method  700 , the packet  708  requests that Station B send a steering matrix as the feedback signal, whereas in the method  200 , the packet  208 , requested a CSI feedback signal. The calibration initiation signal may be sent at a physical layer or a media access control layer, for example, of a wireless communication protocol. The packet  708  may be a sounding packet, for example. 
     At a block  712 , in response to receiving the packet  708 , Station B generates a forward steering matrix, V F , which Station B transmits to Station A via a steering matrix feedback packet  716  at block  718 . The transmission of the steering matrix feedback packet  716  is generally not considered time critical. 
     In response to the steering matrix feedback packet  716 , block  722  determines the partial estimation of the reverse channel H BA  based on the received forward steering matrix, V F , determined by Station B at the block  712 . To determine the reverse channel in this manner, Station A may be configured with the same information Station B uses to determine V F . For example, Station A and Station B may be both designed with the same steering matrix determination schemes. With the determination at the block  722 , after receipt of the pack  716 , Station A will have information on both the forward link, V F , and the reverse link, {tilde over (H)} BA , from which a correction matrix K A,Tx  can be determined. 
     If Station B only uses a portion of its antennas to send V F , block  722  will determine the partial estimation of the reverse channel [{tilde over (H)} BA ] *Ψ     B   . 
     With the estimate of the reverse channel, {tilde over (H)} BA  or [{tilde over (H)} AB ] *Ψ     B   , a block  724  determines the reverse steering matrix, V R , for example applying Equation 17:
 
 V   R   =f ([ {tilde over (H)}   BA   T ] *Ψ     B   ),  (Equ. 17)
         A block  728  then determines the correction matrix based on the determined forward and backward steering matrices, according to Equation 18:
 
 K   A,Tx   =V   F   V   R   H ,  (Equ. 18)
   For even more accurate results, in some examples, the block  728  averages the matrix values over multiple OFDM sub-carriers. Furthermore, as mentioned above, Station A determines the forward steering matrix using the same function, f, that Station B uses in determining the forward steering matrix. Further still, in some examples, Station B uses an identity matrix when determining the forward steering matrix, which simplifies the determination of the reverse steering matrix at Station A.       

     The method  700  of  FIG. 9  is implemented by the system  10  of  FIG. 1 . Of course, in other examples, the method  700  is implemented by other systems. Further, in other examples, the system  10  implements a method other than method  700 . 
     For example, the controller  20  causes the sounding packet  708  to be sent to the receiver  16  (block  704 ). The controller  30  causes the steering matrix feedback packet  716  to be sent to the transmitter  12  (block  718 ). Steering matrix calculation unit  48  generates the forward steering matrix, V F , (block  712 ), and the controller  40  causes this information to be sent back to the transmitter (block  718 ). The correction matrix calculation unit  29  generates [{tilde over (H)} BA ] *Ψ     B    (block  722 ). The steering matrix calculation unit  28  generates the reverse steering matrix, V R , (block  724 ). The correction matrix calculation unit  29  also generates the one or more correction matrices (block  728 ). 
     After initialization, to send a data packet, the space-time mapping block  24  left-multiplies the correction matrix K A,Tx  with the signal to be transmitted at baseband. In other examples, the steering matrix calculation block  28  modifies the steering matrix Q A  using the correction matrix K A,Tx  by, for example, left-multiplying the correction matrix K A,Tx  with the steering matrix Q A  to generate a modified steering matrix. In yet other examples, the correction matrix calculation unit  29  modifies the steering matrix Q A  using the correction matrix K A,Tx . Also, the matrix equalizer  25  left-multiplies the correction matrix K A,Rx  with the signal it receives from the station  16  at baseband. 
     In some examples, Station B uses a single antenna to compute the forward steering matrix, for example using a cophasing technique. Station A, then, may then use a method  800 , similar to that of method  500 , and shown in  FIG. 10 , when Station B is using a single antenna. The feedback packet  716  is always spatially mapped by a 1×1 identity matrix, because there is one transmitted stream from only one antenna at Station B. 
     At block  804 , Station A sends the calibration initiation packet  708  using a single antenna of Station A. At block  806 , Station B (uses 1 receive antenna) determines an estimate of the 1-by-N TX  forward channel [{tilde over (H)} AB ] *i  based on packet  716 . In response, at block  810 , Station B performs a cophasing on the estimate of the 1-by-N TX  forward channel [{tilde over (H)} AB ] *i  to determine the forward steering matrix according to: 
     
       
         
           
             
               
                 
                   
                     
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             At block  812 , Station B sends the forward steering matrix to Station A, which then determines an estimate of the reverse channel [{tilde over (H)} BA   T ] *i  at block  814  before determining the reverse steering matrix at block  818  according to: 
           
         
       
    
     
       
         
           
             
               
                 
                   
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             Station A then returns control to block  728  for determining the correction matrix K A,Tx . 
           
         
       
    
     While the foregoing describes the determination of example correction factors and matrices, in other examples, other correction factors or matrices that compensate for the effects of RF chain impairments and other non-equalities that prevent the partial estimation of the forward channel [{tilde over (H)} AB ] *Ψ     B    from equaling or being the same as the partial estimation of the reverse channel [{tilde over (H)} BA   T ] *Ψ     B    developed from the reverse channel {tilde over (H)} BA  may be determined. These other correction factors or matrices would then be used instead of the correction matrices specifically detailed herein. 
       FIG. 11  is a flow diagram of another example method  900  for calibrating a station in a wireless network. The method  900  will be described with reference to  FIG. 12 , which is a timing diagram illustrating communications between a Station A and a Station B during the calibration method  900 . At a block  904 , Station A initiates a calibration process by sending a packet  908  to Station B. The packet  908  sent at the block  254  may include a “calibration initiation” signal, which indicates to Station B that Station A is requesting to initialize calibration. The calibration initiation signal may be sent at a physical layer or a media access control layer, for example, of a wireless communication protocol. More specifically, the packet  908  includes an acknowledgement request. The packet  908  is not a sounding packet. In other words, the packet  908  is a non-sounding packet. 
     At a block  910 , in response to receiving the packet  908  with the calibration initiation signal, Station B transmits an acknowledgment packet  912  after a time period Δt. The acknowledgment packet  912  may be a non-sounding packet. 
     At a block  916 , in response to receiving the acknowledgment packet  912 , Station A generates an estimate of a potentially partial portion of the reverse channel {tilde over (H)} BA  based on the reception of the acknowledgment packet  912 . 
     At a block  918 , Station A generates and transmits a sounding packet  922  to Station B. The packet  922  is a sounding packet because it includes training information for all available spatial dimensions of a multiple-antenna channel. 
     At a block  924 , in response to receiving the sounding packet  922  with the calibration initiation signal, Station B generates a full-dimensional CSI estimation of {tilde over (H)} AB . At a block  926 , Station B transmits the full-dimensional CSI estimation of {tilde over (H)} AB  back to Station A via a CSI feedback packet  932 . According to many wireless network protocols, transmission of the CSI feedback packet  932  typically is not considered time critical. 
     At a block  934 , after receiving the CSI feedback packet  932 , Station A calculates the correction matrix K A,Tx  using [{tilde over (H)} BA ] *Ψ     B    (block  916 ) and the corresponding columns in {tilde over (H)} AB  (block  924 ) denoted as [{tilde over (H)} AB ] *Ψ     B   . Also, Station A may additionally or alternatively calculate a correction matrix K A,Rx . 
     Although in  FIG. 12  it is shown that Station A transmits the sounding packet  922  after receiving the acknowledgment packet  912 , it should be understood that, in other examples, Station A transmits the sounding packet  922  after receiving or during reception of the acknowledgment packet  912 . 
     In some implementations of the method  900 , Station A requests, via calibration initiation packer  908 , that Station B send the ACK packet  912  using a modulation and coding scheme (MCS) with the same number of spatial streams as the number of antennas in Station A, for example using a link adaptation protocol or another protocol that can request the uplink packet. In implementations of the method  900 , in which Station B uses only a single antenna, then such MCS matching is not used. 
     While the foregoing have been described from the perspective of the transmitter Station A, which is the beamformer station, the techniques may be applied to the receiver Station B, which is the beamformee station. An explicit receiver Station B may be configured to perform calibration by, for example, upon receiving a sounding packet from Station A, requesting a CSI feedback signal or steering matrix feedback signal from Station A. For example, upon receipt of the sounding packet, Station B may send a calibration initiation signal to Station A using a fixed MCS, where Station B forms the calibration initiation signal using the same number of spatial streams as the number of antennas used in the transmitter Station A, so that identity spatial mapping matrix may be used. 
     The implicit beamformer as transmitter Station A may be configured to perform calibration, by sending a sounding packet to the explicit beamformer receiver Station B, requesting either a CSI feedback signal or a steering matrix feedback signal. The Station A, upon receiving the feedback packet, may buffer the channel estimation for this packet, and then process both the feedback (i.e., the CSI feedback signal or the steering matrix) and the reverse link channel estimation upon receiving the feedback packet, K B,Tx , to obtain the calibration coefficients. 
     While the above calibration processes are described, as having functions performed by one or the other of the transmitter and receiver stations, in other examples, each of the functions may be performed by both the transmitter and receiver. That is, in some examples, both an implicit beamforming Station A and an implicit beamforming Station B performs the methods described herein to establish a fully reciprocal communication between the two stations. In the later case, the implicit beamforming Station B acts as an explicit beamformer in calibrating itself, where Station B determines and applies its own correction matrix, whenever transmitting reverse channel packets to Station A. 
     The techniques described above are applicable, for example, in any single-carrier or multi-carrier (e.g., OFDM systems) systems that support data packet transceiving, and with multiple antennas at the transmitter. With a multi-carrier system, calibration processes and calculations such as described above may be conducted for each carrier. For example, in an OFDM system, calibration processes and calculations such as described above may be conducted for each sub-carrier. 
     After the correction matrix K A,Tx  and/or the correction matrix K A,Rx  are generated, these matrices may be stored in the memory  21  or in any other desired memory. The steering matrix calculation unit  28  is, thereafter, simply determining a new steering matrix using implicit beamforming, i.e., by determining an uncompensated or implicit steering matrix using any standard implicit beamforming technique, e.g., based on the assumption that an inferred channel (i.e., the estimate of the forward channel) {tilde over (H)} BA   T  is equal to the actual forward channel {tilde over (H)} AB , but then multiplying the uncompensated steering matrix by the correction matrix K A,Tx  to create a compensated or corrected steering matrix that takes into account the errors introduced by RF chain impairments. As will be understood using this technique, once the correction matrix K A,Tx  has been obtained for the forward channel between a particular transmitter/receiver pair, the transmitter can determine a new steering matrix for the forward channel at any time using implicit beamforming and the correction matrix K A,Tx  (and thus relying only on signals transmitted from the receiver to the transmitter, i.e., in the reverse channel). 
     To illustrate the techniques described herein,  FIG. 13  shows a MIMO communication system  1010  having a single transmitter  1012  with six transmission antennas  1014 A- 1014 F, and a single receiver  1016  with four receiver antennas  1018 A- 1018 D. In this example, the steering matrix is developed by the transmitter  1012  using a corrected steering matrix developed in the manner described above to create a transmit gain pattern  1019  as shown disposed next to the transmitter  1012 . As illustrated in  FIG. 11 , the transmit gain pattern  1019  includes multiple high gain lobes  1019 A- 1019 D generally disposed in the directions of the receiver antennas  1018 A- 1018 D. The high gain lobes  1019 A- 1019 D are orientated in the directions of propagation from the transmitter  1012  to the particular receiver antennas  1018 A- 1018 D while lower gain regions, which may even include one or more nulls, are produced in other directions of propagation. While  FIG. 11  illustrates a separate high gain lobe directed to each of the receiver antennas  1018 A- 1018 D, it will be understood that the actual gain pattern produced by the beam steering matrix calculations using implicit beamforming and a correction matrix may not necessarily include a separate high gain lobe for each of the receiver antennas  1018 A- 1018 D. Instead, the gain pattern developed by the beam steering matrix for the transmitter  1012  may have a single high gain lobe covering or directed generally to more than one of the receiver antennas  1018 A- 1018 D. Thus, it is to be understood that the beam pattern resulting from the creation of a steering matrix using implicit beamforming and a calibration factor may or may not have separate high gain lobes separated by low gain regions or nulls for each of the receiver antennas. 
     Of course, developing the beam pattern  1019  to have high gain regions and low gain regions based on a correction matrix may be performed in any desired manner and location. For example, any of the components within the receiver  16  of  FIG. 1 , including the controller  40 , the steering matrix calculation unit  48  and the channel determination unit  39  determines the CSI or other measured description of the forward channel and, if desired determines the right singular matrixes for the forward channel from this information. The receiver  16  then sends any of this determined information to the transmitter  12 . If desired, however, the receiver  16  simply collects the known signal received from the transmitter  12  and may send this signal back to the transmitter  12  without processing this signal in any significant manner, and the transmitter  12  then determines the measured description of the forward channel from this information. In either case, the controller  20  and/or the steering matrix calculation unit  28  and/or the correction matrix calculation unit  29  within the transmitter  12  uses the information determined about the forward channel and/or the reverse channel to calculate and apply the correction matrix in modifying the steering matrix and/or for use in the space-time mapping block  24  to thereby implement beamforming in the forward channel. 
     It will be understood that the correction matrix equations, e.g., the computation of the correction matrix, may be performed at any desired location within the wireless communication system  10  of  FIG. 1 , including within the controller  20  or other hardware, software, or firmware of the transmitter  12 , as well as within the controller  40  or other hardware, software, or firmware of the receiver  16 . In the later case, the receiver  16  computes at least some of the forward channel information to be used by the transmitter  12  based on the specifics of the forward channel determined at the receiver  16  and, if desired, the CSI developed by the receiver  16 , and sends this information to the transmitter  12  to be used in calculating the correction matrix. On the other hand, the steering matrix for the transmitter space-time mapping block  24  of  FIG. 1  is calculated by the steering matrix calculation unit  28  within the transmitter  12  based on raw channel data or signals sent by the receiver  16  provided and sent back from the receiver  16  to the transmitter  12 , in other examples. 
     Of course, the beamforming technique described herein is not limited to being used in a transmitter of a MIMO communication system communicating with a single receiver of the MIMO communication system, but can additionally be applied when a transmitter of a MIMO communication system is communicating with multiple receivers, each of which has one or more receiver antennas associated therewith. In this case, the transmitter performs or implement a separate correction matrix calculation for each receiver to which the transmitter will transmit and may therefore develop a different steering matrix and/or correction matrix for each of the possible receivers, and uses those steering matrixes to beamform to the separate or different receivers at different times or using different channels, e.g., OFDM channels, of the system. Moreover, while the maximum gains of the high gain lobes of each of the transmit gain patterns illustrated in  FIG. 11  are shown as being the same, the steering matrix calculation units  28  and  48  may develop steering matrixes which produce high gain lobes with differing maximum gains. 
     While the beamforming and correction matrix calculations described herein are described in one example as being implemented in hardware, these calculations alternatively or additionally be implemented in software stored in, for example, one of the memories  21 ,  41  and implemented on a processor associated with one or both of the controllers  20 ,  40 , the steering matrix calculation units  28 ,  48  and/or the units  29  and  39  of the MIMO communication system  10  of  FIG. 1 , or implanted in firmware as desired. If implemented in software, the routines may be stored in any computer readable memory such as in RAM, ROM, flash memory, a magnetic disk, a laser disk, or other storage medium. Likewise, this software may be delivered to a MIMO system device (such as a transmitter or a receiver) via any known or desired delivery method including, for example, over a communication channel such as a telephone line, the Internet, a wireless connection, etc., or via a transportable medium, such as a computer-readable disk, flash drive, etc. 
     More generally, the various blocks, operations, and techniques described above may be implemented in hardware, firmware, software, or any combination of hardware, firmware, and/or software. When implemented in hardware, some or all of the blocks, operations, techniques, etc. may be implemented in, for example, a custom integrated circuit (IC), an application specific integrated circuit (ASIC), a field programmable logic array (FPGA), a programmable logic array (PLA), etc. 
     When implemented in software, the software may be stored in any computer readable memory such as on a magnetic disk, an optical disk, or other storage medium, in a RAM or ROM or flash memory of a computer, processor, hard disk drive, optical disk drive, tape drive, etc. Likewise, the software may be delivered to a user or a system via any known or desired delivery method including, for example, on a computer readable disk or other transportable computer storage mechanism or via communication media. Communication media typically embodies computer readable instructions, data structures, program modules or other data in a modulated data signal such as a carrier wave or other transport mechanism. The term “modulated data signal” means a signal that has one or more of its characteristics set or changed in such a manner as to encode information in the signal. By way of example, and not limitation, communication media includes wired media such as a wired network or direct-wired connection, and wireless media such as acoustic, radio frequency, infrared and other wireless media. Thus, the software may be delivered to a user or a system via a communication channel such as a telephone line, a DSL line, a cable television line, a wireless communication channel, the Internet, etc. (which are viewed as being the same as or interchangeable with providing such software via a transportable storage medium). 
     The present invention may be embodied in any type of wireless communication system including, for example, ones used in wireless computer systems such as those implemented via a local area network or a wide area network, internet, cable and satellite based communication systems (such as internet, data, video and voice communication systems), wireless telephone systems (including cellular phone systems, voice over internet protocol (VoIP) systems, home-based wireless telephone systems, etc.) 
     Moreover, while the present invention has been described with reference to specific examples, which are intended to be illustrative only and not to be limiting of the invention, it will be apparent to those of ordinary skill in the art that changes, additions and/or deletions may be made to the disclosed embodiments without departing from the spirit and scope of the invention.