Patent Publication Number: US-9887535-B2

Title: Short circuit protection for a power switch

Description:
TECHNICAL FIELD 
     The present document relates to the protection of a power switch, e.g. of the pass device of a voltage regulator, subject to a short circuit at an output port of the power switch. 
     BACKGROUND 
     An electronic circuit which provides electrical power to a load, e.g. a voltage regulator such as a low-dropout (LDO) regulator, may comprise a bypass switch in order to provide a direct link to the supply voltage VDD of the electronic circuit. Such a direct link may be beneficial for increasing the reaction speed of the electronic circuit subject to a load transient. The bypass switch may be in a situation where, due to external conditions or due to nonlinear conditions such as a bypass mode, a regulation device of the electronic circuit (e.g. an intermediate amplification stage) is fully opened in order to try to regulate the output voltage of the electronic circuit. As such, the bypass switch may be in a state of maximum current capability also known as a state of lowest resistance, regardless of the actual loading situation of the electronic circuit. 
     If an external event causes a “short circuit”, which is a low-ohmic low inductive connection to another node, while the bypass switch is in a state of lowest resistance, a situation may occur where a significant current is drawn directly from the supply voltage VDD of the electronic circuit. It is desirable to limit or to interrupt such a short circuit current through the bypass switch, as the bypass switch and/or a power supply which provides the supply voltage may be damaged by substantial short circuit currents. 
     SUMMARY 
     The present document addresses the technical problem of providing efficient and reliable means for reducing the current through a power switch or power transistor subject to a short circuit situation. According to an aspect, a power providing circuit which is configured to provide a current at an output voltage to a load at an output of the power providing circuit is described. The power providing circuit may be or may comprise bypass circuitry, e.g. bypass circuitry for a voltage regulator. Alternatively or in addition, the power providing circuit may comprise or may be a voltage regulator (e.g. a low dropout regulator). 
     The power providing circuit comprises a power transistor (which may also be referred to as a power switch, a bypass transistor or a bypass switch) which is configured to draw the current for the load from a supply voltage. The power transistor may comprise a bipolar junction transistor (BJT), an insulated-gate bipolar transistor (IGBT) and/or a metal oxide semiconductor (MOS) transistor. In general terms, the power transistor may comprise any kind of electrical controllable switch. A resistance (e.g. a drain-source resistance in case of a MOS transistor) of the power transistor is controlled using a control voltage (e.g. a gate voltage in case of a MOS transistor) which is applied to a control port (e.g. a gate in case of a MOS transistor) of the power transistor. By varying the resistance, the current through the power transistor may be controlled, i.e. the current through the power transistor which is drawn from the supply voltage may be controlled via the control voltage. 
     The power transistor may comprise a p-type metal oxide semiconductor (PMOS) transistor. The source of the power transistor may be (directly) coupled to the supply voltage and the drain of the power transistor may be (directly) coupled to the output of the power providing circuit. In more general terms, the source may be referred to as a first port and the drain may be referred to as a second port. Hence, the first port of the power transistor may be (directly) coupled to the supply voltage and the second port of the power transistor may be (directly) coupled to the output of the power providing circuit. As such, the current through the power transistor may be directly drawn from the supply voltage and may be directly provided to the load. 
     The power providing circuit further comprises short circuit protection circuitry which is configured to couple the control port of the power transistor with the first port of the power transistor to put the power transistor in an off-state, subject to a drop of the output voltage. The short circuit protection circuitry comprises a short circuit control transistor which comprises a second port (e.g. a drain) and a first port (e.g. a source) that are configured to couple the control port (e.g. the gate) and the first port (e.g. the source) of the power transistor with one another. In particular, the short circuit control transistor may be configured to provide a direct connection between the control port and the first port of the power transistor, via a conduction channel (e.g. via the drain-source channel) of the short circuit control transistor (without any further intermediate components). A state and/or the resistance (notably the drain-source resistance) of the short circuit control transistor may be controlled via the control port (e.g. the gate) of the short circuit control transistor, and a voltage level at the control port (e.g. the gate) of the short circuit control transistor may (e.g. only) depend on the output voltage. In particular, the voltage level at the control port (e.g. the gate) of the short circuit control transistor may be independent of a current limit of current limit control circuitry of the power providing circuit and/or of a reference voltage which may be used to set a level of the output voltage of the power providing circuit. 
     In a similar manner to the power transistor, the short circuit control transistor may comprise or may be a p-type metal oxide semiconductor transistor. The drain of the short circuit control transistor may be (directly) coupled to the gate of the power transistor and the source of the short circuit control transistor may be (directly) coupled to the source of the power transistor. In more general terms, the second port of the short circuit control transistor may be (directly) coupled to the control port of the power transistor and the first port of the short circuit control transistor may be (directly) coupled to the first port of the power transistor. 
     By making use of short circuit protection circuitry which is configured to provide a short circuit between the control port and the first port of the power transistor directly in response to a drop of the output voltage, the current through the power transistor may be interrupted rapidly in response to a “short circuit” condition at the output of the power providing circuit. As such, efficient, reliable and fast means for reducing the current through a power transistor subject to a short circuit situation are provided. 
     The short circuit protection circuitry (in particular, the short circuit control transistor) may be configured to couple the control port of the power transistor with the first port of the power transistor only if a “short circuit” condition is met. Otherwise, the short circuit protection circuitry (in particular, the short circuit control transistor) may not couple the control port to the first port of the power transistor. As such, the short circuit protection circuitry may be inactive during normal operation of the power providing circuit, and may only be used in case a “short circuit” condition is met. By doing this, the short circuit protection circuitry may be added to the power providing circuit without impacting other components of the power providing circuit during normal operation of the power providing circuit (e.g. a regulation loop of current limit circuitry and/or a regulation loop of a voltage regulator). 
     The “short circuit” condition may be met if a slope of the drop of the output voltage exceeds a pre-determined slope-threshold and/or if a level of the drop of the output voltage exceeds a pre-determined level-threshold. The short circuit protection circuitry (in particular, the short circuit control transistor) may be designed in accordance to the slope-threshold and/or the level-threshold. In particular, a threshold voltage of the short circuit control transistor may be selected in dependence of the slope-threshold and/or the level-threshold. Alternatively or in addition, a driver circuit of the short circuit control transistor (which is configured to derive the voltage level at the gate of the short circuit control transistor based on the output voltage) may be dependent on the slope-threshold and/or the level-threshold. By doing this, it may be ensured that the short circuit control transistor is only activated if the “short circuit” condition is met, and that the short circuit control transistor remains open if the “short circuit” condition is not met (e.g. because the slope and/or the level of the drop of the output voltage are not sufficiently high). 
     The short circuit protection circuitry (in particular the driver circuit for the short circuit control transistor) may comprise high pass filtering means which are configured to derive the voltage level at the control port (e.g. the gate) of the short circuit control transistor by high pass filtering the output voltage. The high pass filtering means may be designed in dependence of the slope-threshold. In particular, the high pass filtering means may be configured to modify the voltage level at the control port (e.g. the gate) of the short circuit control transistor such that the short circuit control transistor goes from an off-state to an on-state only if the “short circuit” condition is met. In other words, the high pass filtering means may be configured to provide a voltage level at the control port of the short circuit control transistor such that the threshold voltage of the short circuit control transistor is exceeded, if the short circuit condition is met. Furthermore, the high pass filtering means may be configured to, otherwise, provide a voltage level at the control port of the short circuit control transistor such that the threshold voltage of the short circuit control transistor is not exceeded (in order to ensure that the short circuit control transistor remains open, whenever the “short circuit” condition is not met). 
     The short circuit protection circuitry (in particular the driver circuit of the short circuit control transistor, and even more particularly the high pass filtering means) may comprise a filtering resistor which is (directly) coupled between the control port and the first port of the short circuit control transistor. Furthermore, the short circuit protection circuitry (in particular the driver circuit of the short circuit control transistor, and even more particularly the high pass filtering means) may comprise a filtering capacitor which is (directly) coupled to the control port of the short circuit control transistor at one side and to the output of the power providing circuit at another side of the filtering capacitor. The filtering resistor and the filtering capacitor may form an RC circuit with a time constant. The time constant may be dependent on the slope-threshold and/or the level-threshold. Such an RC circuit provides efficient means for driving the short circuit control transistor such that the short circuit control transistor is only activated if the “short circuit” condition is met. 
     The short circuit protection circuitry (in particular the driver circuit of the short circuit control transistor, and even more particularly the high pass filtering means) may comprise an amplifier and/or attenuator which is configured to amplify and/or attenuate a voltage which is derived from the output voltage, e.g. a voltage at the output of the high pass filtering means, prior to applying the attenuated/amplified voltage to the gate of the short circuit control transistor. In particular, the voltage at a midpoint between the filtering resistor and the filtering capacitor may be amplified/attenuated by the amplifier/attenuator. The output of the amplifier/attenuator may be (directly) coupled to the control port of the short circuit control transistor. The gain of the amplifier/attenuator may depend on the threshold voltage of the short circuit control transistor and/or on the level-threshold. As such, the amplifier/attenuator may be used to tune the short circuit protection circuitry to be activated only if the “short circuit” condition is met. 
     The power providing circuit may further comprise current limit circuitry which is configured to limit the current through the power transistor in accordance to a pre-determined current limit. The current limit circuitry may comprise sensing means which are configured to provide an indication of the current through the power transistor. Furthermore, the current limit circuitry may comprise comparing means which are configured to provide a feedback voltage by comparing the indication of the load current with the pre-determined current limit. In addition, the current limit circuitry may comprise feedback means which are configured to set the control voltage (e.g. the gate voltage) at the control port of the power transistor in dependence of the feedback voltage. 
     As such, the current limit circuitry may be used to limit the current through the power transistor and to thereby protect the power transistor and the supply voltage (during normal operation of the power providing circuit). However, the short circuit protection circuitry may be configured to couple the control port with the first port of the power transistor within a first reaction time interval, subject to the drop of the output voltage, and the current limit circuitry may be configured to limit the current through the power transistor in accordance to the pre-determined current limit within a second reaction time interval, subject to the drop of the output voltage. The first reaction time interval may be smaller than the second reaction time interval. The relatively fast reaction time interval of the short circuit protection circuitry may be achieved by the direct coupling of the control port and the first port of the power transistor using the short circuit control transistor. On the other hand, the current limit circuitry typically comprises a feedback loop which allows for the setting and/or regulation of the current limit. Such a feedback loop may be relatively slow compared to the direct coupling of the control port and the first port of the power transistor which is achieved by the short circuit control transistor. 
     As such, the power providing circuit may comprise a combination of relatively slow and precise current limit circuitry (for setting a current limit during “normal” operation of the power providing circuit, i.e. in cases when the “short circuit” condition is not met) and short circuit projection circuitry (for interrupting the current through the power transistor in short circuit situations, i.e. in cases when the “short circuit” condition is met). 
     The sensing means of the current limit circuitry may comprise a sensing transistor having a control port (e.g. a gate) that is (directly) coupled to the control port of the power transistor, and having a first port (e.g. a source) that is (directly) coupled to the first port of the power transistor, and having a second port (e.g. a drain) that is coupled to the second port (e.g. the drain) of the power transistor via a current mirror. Furthermore, the feedback means may comprise a feedback transistor having a control port (e.g. a gate) that the feedback voltage is applied to, wherein the control voltage is dependent on a resistance (e.g. a drain-source resistance) of the feedback transistor. In addition, the comparing means may comprise a current mirror which is configured to map a current through the sensing transistor to the control port (e.g. the gate) of the feedback transistor, and a current source which is configured to provide a current to the control port (e.g. the gate) of the feedback transistor which is in accordance to the current limit. 
     As indicated above, the power providing circuit may comprise or may be a voltage regulator which is configured to regulate the output voltage in accordance to a reference voltage. The voltage regulator may comprise voltage sensing means which are configured to provide an indication of the output voltage, and a differential amplification stage which is configured to provide an input voltage, based on the reference voltage and based on the indication of the output voltage at the output node. The control voltage which is applied to the control port of the power transistor may depend on the input voltage (e.g. via an intermediate or second amplification stage of the voltage regulator). 
     As such, the control of the power transistor may be embedded into the regulation of a voltage regulator. In the context of such a regulation, it may occur that the power transistor is set to have a relatively low resistance, notably drain-source resistance (e.g. subject to a load transient of the load at the output of the power providing circuit). The short circuit projection circuitry described in the present document is particularly beneficial to protect the power transistor and/or the supply voltage if a short circuit situation occurs when the power providing circuit is operated in such a situation. 
     The voltage regulator may comprise a pass device which is configured to provide a current to the load. The pass device may be arranged in parallel to the power transistor between the supply voltage and the output of the power providing circuit. As such, the power transistor may assist the pass device in providing additional current to the output of the power providing circuit in case of load transients. 
     According to another aspect, a method for protecting a power transistor in case of a short circuit situation is described. The method comprises drawing current for a load from a supply voltage via a power transistor, wherein a resistance (e.g. a drain-source resistance) of the power transistor is controlled using a control voltage (e.g. a gate voltage) which is applied to a control port (e.g. a gate) of the power transistor and wherein the current is provided to the load at an output voltage. Furthermore, the method comprises, subject to a drop of the output voltage, coupling the control port of the power transistor with a first port (e.g. a source) of the power transistor, to put the power transistor in an off-state. 
     It should be noted that the methods and systems including its preferred embodiments as outlined in the present document may be used stand-alone or in combination with the other methods and systems disclosed in this document. In addition, the features outlined in the context of a system are also applicable to a corresponding method. Furthermore, all aspects of the methods and systems outlined in the present document may be arbitrarily combined. In particular, the features of the claims may be combined with one another in an arbitrary manner. 
     In the present document, the term “couple” or “coupled” refers to elements being in electrical communication with each other, whether directly connected e.g., via wires, or in some other manner. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention is explained below in an exemplary manner with reference to the accompanying drawings, wherein 
         FIG. 1 a    illustrates an example block diagram of an LDO regulator; 
         FIG. 1 b    illustrates the example block diagram of an LDO regulator in more detail; 
         FIG. 2  shows a circuit diagram of an example bypass switch with a current limit circuit; 
         FIGS. 3 a , 3 b  and 3 c    show circuit diagrams of example short circuit protection circuitry; 
         FIGS. 4 a  and 4 b    show circuit diagrams of example short circuit protection circuitry for a bipolar transistor; and 
         FIG. 5  shows a flow chart of an example method for protecting a power switch subject to a short circuit. 
     
    
    
     DESCRIPTION 
     As outlined above, electronic power providing circuits such as voltage regulators may comprise one or more bypass switches which are configured to couple an output of the power providing circuit directly with a supply voltage of the power providing circuit. This may be beneficial for increasing the reaction speed of the power providing circuit subject to load transients at the output of the power providing circuit. An example of a power providing circuit is an LDO regulator. A typical LDO regulator  100  is illustrated in  FIG. 1 a   . The LDO regulator  100  comprises an output amplification stage  103 , e.g. a field-effect transistor (FET), at the output and a differential amplification stage or differential amplifier  101  (also referred to as error amplifier) at the input. A first input (fb)  107  of the differential amplifier  101  receives a fraction of the output voltage V out  determined by the voltage divider  104  comprising resistors R 0  and R 1 . The second input (ref) to the differential amplifier  101  is a stable voltage reference V ref    108  (also referred to as the bandgap reference). If the output voltage V out  changes relative to the reference voltage V ref , the drive voltage to the output amplification stage, e.g. the power FET, changes by a feedback mechanism called main feedback loop to maintain a constant output voltage V out . 
     The LDO regulator  100  of  FIG. 1 a    further comprises an additional intermediate amplification stage  102  configured to amplify the output voltage of the differential amplification stage  101 . As such, an intermediate amplification stage  102  may be used to provide an additional gain within the amplification path. Furthermore, the intermediate amplification stage  102  may provide a phase inversion. 
     In addition, the LDO regulator  100  may comprise an output capacitance C out  (also referred to as output capacitor or stabilization capacitor or bybass capacitor)  105  parallel to the load  106 . The output capacitor  105  is used to stabilize the output voltage V out  subject to a change of the load  106 , in particular subject to a change of the load current I load . It should be noted that typically the output current I out  at the output of the output amplification stage  103  corresponds to the load current I load  through the load  106  of the regulator  100  (apart from typically minor currents through the voltage divider  104  and the output capacitance  105 ). Consequently, the terms output current I out  and load current I load  are used synonymously, if not specified otherwise. 
     Typically, it is desirable to provide a stable output voltage V out , even subject to (positive or negative) transients of the load  106 . By way of example, the regulator  100  may be used to provide a stable output voltage V out  to the processor of an electronic device (such as a smartphone). The load current I load  may vary significantly between a sleep state and an active state of the processor, thereby varying the load  106  of the regulator  100 . In order to ensure a reliable operation of the processor, the output voltage V out  should remain stable, even in response to such load transients. In particular, overvoltage and/or undervoltage situations of the output voltage V out  should be avoided. 
     At the same time, the LDO regulator  100  should be able to react rapidly to load transients, i.e. the LDO regulator  100  should be able to rapidly provide the requested load current I load , subject to a load transient. This means that the LDO regulator  100  should exhibit a high bandwidth. 
       FIG. 1 b    illustrates the block diagram of a LDO regulator  120 , wherein the output amplification stage A 3  (reference numeral  103 ) is depicted in more detail. In particular, the pass transistor or pass device  201  and the driver stage  110  of the output amplification stage  103  are shown. Typical parameters of an LDO regulator are a supply voltage of 3V, an output voltage of 2V, and an output current or load current ranging from 1 mA to 100 or 200 mA. Other configurations are possible. The present invention is described in the context of a linear regulator. It should be noted, however, that the present invention is applicable to power provisioning circuits in general. 
     As indicated above, a bypass circuit may be provided in conjunction with a voltage regulator  100 ,  120  in order to increase the reaction speed of the voltage regulator  100 ,  120  subject to a load transient. An example bypass circuit  300  is illustrated in  FIG. 2 . The bypass circuit may be considered to be an example of a generic power providing circuit. The bypass circuit  300  comprises a power transistor  301  (also referred to herein as a bypass transistor) which is configured to couple the output voltage V out    321  with the supply voltage VDD  322  (also referred to as Vin in  FIG. 1 b   ). As such, the power transistor  301  may be arranged in parallel to the pass device  201  of the voltage regulator  100 ,  120 . In another example arrangement, the power transistor  301  corresponds to (e.g. is equal to) the pass device  201  of the voltage regulator  100 ,  120 . In the illustrated example in  FIG. 2 , the power transistor  301  is a PMOS transistor. 
     The power transistor  301  may be controlled by the gate voltage  325  which is applied to the gate of the power transistor  301 . The gate voltage  325  is set using the input control transistor  311 , a state of the input control transistor  311  being dependent on the enabling voltage  323  which is applied to the gate of the input control transistor  311 . The enabling voltage  323  may correspond to (e.g. be equal to) the output voltage of the second amplification stage  102  (i.e. out_s2) of the voltage regulator  100 ,  120  shown in  FIGS. 1 a , 1 b   . In the illustrated example, the input control transistor  311  is a NMOS transistor. 
     If the input control transistor  311  is enabled using the enabling voltage  323 , the gate voltage  325  is pulled down, thereby enabling the power transistor  301 . A level of the gate voltage  325  may be set by a resistance value of the gate resistor  312 . As such, the power transistor  301  may be opened in dependence of the level of the enabling voltage  323 , thereby providing a current to the output node of the bypass circuit  300  and thereby stabilizing the output voltage  321 . 
     The bypass circuit  300  of  FIG. 2  further comprises current limit circuitry which is configured to limit the current through the power transistor  301 . The current limit circuitry comprises a sensing transistor  304  which is configured to sense a state of the power transistor  301 . The sensing transistor  304  may correspond to the power transistor  301 , however, at reduced dimensions. In particular, the sensing transistor  304  may be configured to provide an indication of the current through the power switch. In other words, the current through the sensing transistor  304  may correspond to (e.g. may be equal to) the current through the power transistor  301 , by a pre-determined scaling factor, wherein the scaling factor depends on the dimensions of the sensing transistor  304  in relation of the dimensions of the power transistor  301 . 
     The current limit circuitry may further comprise a first current mirror comprising the transistors  302 ,  305 . The current mirror is coupled to the drain of the power transistor  301  (which corresponds to the output node of the bypass circuit  300 ) and to the drain of the sensing transistor  304 . By doing this, the output voltage  321  may be “copied” from the drain of the power transistor  301  to the drain of the sensing transistor  304 , thereby setting the sensing transistor  304  to the same operating point as the power transistor  301 . The current through the sensing transistor  304  (which is also referred to as the sensing current) provides an indication of the current through the power transistor  301  (e.g. is proportional to the current through the power transistor  301 ). 
     The sensing current may be compared with a pre-determined current limit which is set by a first current source  308 . For this purpose, a second current mirror which comprises the transistors  306 ,  307  may be used as illustrated in  FIG. 2 . The second current mirror “copies” the sensing current from the transistor  306  to the transistor  307  (possibly in an amplified or attenuated manner). A drain of the transistor  307  is coupled to the first current source  308 , such that the sensing current is compared to the current limit. The drain of the transistor  307  is coupled to the gate of a feedback transistor  310 , which is implemented as a NMOS transistor in the illustrated example. If the current limit is higher than the (amplified/attenuated) sensing current, then the gate of the feedback transistor  310  is pulled up, thereby opening the feedback transistor  310  and thereby allowing the power transistor  301  to be controlled by the enabling voltage  323 . On the other hand, if the current limit is smaller than the (amplified/attenuated) sensing current, then the gate of the feedback transistor  310  is pulled down, thereby closing the feedback transistor  310  and thereby interrupting the current flowing through the enabling transistor  311  (regardless the level of the enabling voltage  323 ). As a result of this, the gate voltage  325  is pulled up, thereby closing the power transistor  301 , and thereby reducing the current through the power transistor  301 . 
     As such, the current limit circuitry provides a regulation loop for ensuring that the current through the power transistor  301  does not exceed a level which is set using the current limit that is provided by the first current source  308 . 
     The current limit circuitry may further comprise a gate capacitor  309  which couples the gate of the feedback transistor  310  to ground  324 . Such a gate capacitor  309  may be used to ensure a reliable deactivation of the current limit, in case of a situation where the current through the power transistor  301  does not exceed the current limit. Furthermore, a second current source  303  may be used to compensate the drain-source voltage across the transistor  306 , in order to ensure a precise alignment of the operating points of the power transistor  301  and of the sensing transistor  304 . 
     Due to the design of the current limit circuitry and in order to keep the current consumption relatively low, the activation time of the current limit circuitry may be relatively long prior to limiting the current through the power transistor  301 . This may lead to a situation, notably in case of a short circuit, where the current through the power transistor  301  becomes very high, thereby overloading the supply voltage  322 . Due to the low-ohmic nature of a short circuit, the spread and speed of the peak current through the power transistor  301 , subject to a short circuit, may be highly variable. 
       FIG. 3 a    illustrates example short circuit protection circuitry  400  which may be used to protect the supply voltage  322  and the power transistor  301  from a short circuit situation. The short circuit protection circuitry  400  is preferably used in conjunction with current limit circuitry (as the one illustrated in  FIG. 2 ). The short circuit protection circuitry comprises a short circuit control transistor  403  which is configured to couple the gate of the power transistor  301  with the supply voltage  322  in order to close the power transistor  301 . In the illustrated example, the short circuit control transistor  403  is implemented as a PMOS transistor. 
     The short circuit control transistor  403  is controlled based on the output voltage  321 . As a result of a short circuit (simulated using the switch  410 ), the output voltage  321  typically exhibits a substantial drop which leads to a drop of the voltage at the gate of the short circuit control transistor  403 , thereby opening the short circuit control transistor  403  and thereby coupling the gate of the power transistor  301  to the supply voltage  322 . This causes the power transistor  301  to close, thereby reducing the current which is drawn from the supply voltage  322 . 
     The short circuit protection circuitry may further comprise means for filtering the output voltage  321  (also referred to herein as high pass filtering means). In particular, the means for filtering may be configured as a high pass filter, such that only relatively fast variations (such as a drop which is caused by a short circuit) are passed to the gate of the short circuit control transistor  403 , and such that relatively slow variations of the output voltage  321  are not passed to the gate of the short circuit control transistor  403 . As a result of this, the gate of the short circuit control transistor  403  may float (in the absence of a substantial drop of the output voltage  321 ) such that the short circuit control transistor  403  remains open, thereby not coupling the gate of the power transistor  301  from the drain of the power transistor  301 . 
     The means for filtering the output voltage  321  may comprise an RC circuit  401 ,  402  as illustrated in  FIG. 3 a   , wherein the RC circuit  401 ,  402  comprises a filtering resistor  402  and a filtering capacitor  401 . The filtering resistor  402  is arranged between the gate and the source of the short circuit control transistor  403  and the filtering capacitor  401  is used to couple the gate of the short circuit control transistor  403  with the output voltage  321 . As a result of this, a rapid transient of the output voltage  321  is directly coupled to the gate of the short circuit control transistor  403 . In particular, a drop of the output voltage  321  directly affects the voltage at the gate of the short circuit control transistor  403 , thereby closing the short circuit control transistor  403 . On the other hand, relatively slow variations of the output voltage  321  are filtered by the filtering capacitor  401  and the filtering resistor  402 , such that the state of the short circuit control transistor  403  remains unaffected (i.e. open) by such slow variations. 
     The RC circuit  401 ,  402  has the additional affect that the voltage at the gate of the short circuit control transistor  403  re-increases with increasing time interval since the drop of the output voltage  321 . In particular, the gate of the short circuit control transistor  403  goes back into a floating state, thereby re-opening the short circuit control transistor  403 , and thereby re-opening the power transistor  301 . The length of the time interval between the closing of the short circuit control transistor  403  and the re-opening of the short circuit control transistor  403  depends on the time constant τ=RC of the RC circuit  401 ,  402 , wherein R is the resistance of the filtering resistor  402  and wherein C is the capacitance of the filtering capacitor  401 . The time constant τ may be selected to be sufficiently large to allow the current limit circuitry to react to the short circuit situation. In other words, the short circuit protection circuitry  400  may be configured to automatically disable itself after a pre-determined time interval following a short circuit, thereby bridging a reaction time of current limit circuitry of the power provisioning circuit  300 . 
     Hence, rather than waiting for the current limit regulation of the current limit circuitry to become active, this process is short-cut by switching the power transistor  301  from a conduction-mode into an off-mode. As a result of this, the current which is drawn from the supply voltage  322  is stopped. 
     In order to allow for a stable current regulation of the current limit circuitry the frequency response of the current limit circuitry typically needs to be limited, thereby creating a delay in the reaction of the current limit circuitry to a short circuit. Furthermore, due to the low power nature of the current limit circuitry the limiting reaction of the current limit circuitry may take a certain time interval until the power transistor  301  is turned off. During this time interval a substantial current is flowing through the power transistor  301 , wherein the current is only limited by a very small on-resistance (i.e. drain-source resistance) of the power transistor  301 . 
     The short circuit protection circuitry  400  is configured to observe the output voltage  321 . Once a fast drop of the output voltage  321  is detected (with a relatively large slope having a magnitude that is larger than a slope-threshold), the short circuit control transistor  403  is activated to discharge the gate of the power transistor  301  quickly which stops any current through the power transistor  301 . 
     Subsequently, the current limit circuitry re-activates the power transistor  301  coming from an off-state of the power transistor  301  (and not coming from an overdriven on-state of the power transistor  301 ). This ensures a stable activation of the regulation mode of the current limit circuitry. 
     As already indicated above, the current limit circuitry of  FIG. 2  requires a certain time interval prior to limiting the current through the power transistor  301 . As a result of this, a relatively large current may be drawn from the supply voltage  322  prior to an activation of the current limit. This leads to a situation where a trade-off needs to be made between a low switch resistance of the power transistor  301 , a limited short circuit current and a low internal current consumption of the current limit circuitry. 
     In order to address this technical problem, an RC element  401 ,  402  which is connected to the output voltage  321  may be used to detect a fast drop of the output voltage  312  which is the indication of a “short circuit” condition. Such a “short circuit” condition may be distinguished from a “load transient” condition by a level of the drop of the output voltage  321 . Typically the level of the voltage drop in case of a “short circuit” condition is higher than the level of the voltage drop in case of a “load transient” condition. 
     The level of the voltage drop of a “short circuit” condition may correspond to the threshold voltage of the short circuit control transistor  403 . As such, the threshold voltage of the short circuit control transistor  403  may be set such that the short circuit control transistor  403  is only closed, if the level of the voltage drop is sufficiently high to indicate a “short circuit” condition (in contrast to a “load transient” condition). As shown in  FIG. 3 b    an amplifier  421  may be used to adjust the appropriate level of voltage drop to the threshold voltage of the short circuit control transistor  403 . Alternatively of in addition, a differential input stage may be used to activate the short circuit control transistor  403  if the voltage drop of the output voltage  321  is lower than the threshold voltage of the short circuit control transistor  403 . This, however, may increase the current consumption for ensuring a fast operation of the short circuit protection circuitry  400 . 
       FIG. 3 b    shows example short circuit protection circuitry  400  which comprises a gate voltage amplifier  421 . The gate voltage amplifier  421  may be used to adjust the reaction speed of the short circuit protection circuitry  400  by amplifying the voltage at the midpoint between the filtering capacitor  401  and the filtering resistor  402  and by applying the amplified voltage to the gate of the short circuit control transistor  403 . As such, an amplifier  421  may be used to maximally short the gate of the power transistor  301 . The stop of current flow from the supply voltage  322  typically increases the slope of the discharge. This forms a positive feedback loop until the output is discharged and the slope zeros. 
     While the short circuit projection circuitry  400  is activated, notably while the short circuit control transistor  403  couples the gate to the source of the power transistor  301 , other circuitry (e.g. current limits) of the power providing circuit  300  may be turned off temporarily, in order to prevent adverse effects of the gate shorting either by function and/or capacitive coupling effects. 
     The short circuit projection circuitry  400  only makes use of the drop of the output voltage  321  as an indication for a “short circuit” condition. The drop of the output voltage  321  typically requires a certain current flow through the power transistor  301 . The level of the current flowing through the power transistor  301  typically depends on the resistance of the power transistor  301 . As such, the resistance of the power transistor  301  influences the distinction between a “short circuit” condition and a “load transient” condition. 
     Typically the peak current through the power transistor  301  increases with the delay in the detection of the “short circuit” condition and in the activation of the short circuit protection circuitry  400 . 
       FIG. 3 c    shows example short circuit protection circuitry  400  which comprises a current source  432  which is referred to herein as the filtering current source  432 . In the illustrated example, the filtering current source  432  replaces the filtering resistor  402 . The filtering current source  432  may be used for detecting the slope of a drop of the output voltage  321 . In particular, the filtering current source  432  may be configured to provide a pre-determined current, wherein the pre-determined current allows to tune the short circuit protection circuitry  400  such that the short circuit control transistor  403  is closed if the slope of the drop of the output voltage  321  is indicative of a “short circuit” condition, and such that the short circuit control transistor  403  is left open if the slope of the drop of the output voltage  321  is indicative of a “load transient” condition. As such, the pre-determined current which is provided by the filtering current source  432  may depend on the slope-threshold. 
     It can be shown experimentally how the load current at the output of the power providing circuit  300  increases in reaction to a short circuit. As a result of this, the output voltage  321  drops. The drop of the output voltage  321  triggers the short circuit control transistor  403  to switch off the power transistor  301 . This is achieved by pulling the gate voltage  325  at the gate of the power transistor  301  high. After a time interval (which depends on the time constant of the RC circuit  401 ,  402 ) the short circuit control transistor  403  opens again such that the gate of the power transistor  301  is released again. As a result of this, the relatively slower current regulation starts up and regulates the output current of the power transistor  301  to the target value given by the current limit which is set by the first current source  308  (if such regulation is possible due to external conditions). A return to normal operation is possible as soon as the short circuit condition is removed. 
     As such, it may be observed that the current increase through the power transistor  301 , subject to a short circuit, may be limited using the short circuit protection circuitry  400  which is described in the present document. A change of slope of the output voltage  321  may be observed during the activation of the short circuit protection circuitry  400 . Such a change of slope indicates that the current increase from the supply voltage  322  has been stopped within the transition of the slope of the output voltage  321 . 
     It should be noted that the short circuit protection circuitry  400  may also be applied to a bipolar junction transistor (BJT) and/or an insulated-gate bipolar transistor (IGBT). Furthermore, one or more of the transistors of the short circuit protection circuitry  400  may comprise or may be implemented as bipolar junction transistors (BJT) and/or insulated-gate bipolar transistors (IGBT).  FIGS. 4 a  and 4 b    show short circuit protection circuitry  400  in conjunction with a bipolar junction transistor  451 . The bipolar transistor  451  is coupled to the short circuit protection circuitry  400  in an analogous manner as the power transistor (MOSFET)  301 , wherein the gate of the power transistor  301  corresponds to the base of the BJT  451 , the source of the power transistor  301  corresponds to the emitter of the BJT  451  and the drain of the power transistor  301  corresponds to the collector of the BJT  451 . In the present document, the MOS transistor  301  and the BJT  451  are referred to as power transistors. Furthermore, the gate/base is referred to as the control port, the source/emitter is referred to as a first port and the drain/collector is referred to as a second port. Furthermore, the resistance of a power transistor  301 ,  451  corresponds to the drain-source resistance of a MOS transistor  301  or to the collector-emitter resistance of a BJT  451 . 
       FIG. 5  shows a flow chart of an example method  500  for protecting a power transistor  301  in case of a short circuit situation at an output of a power providing circuit  300 . The method  500  comprises drawing  501  current for a load  106  from a supply voltage  322  via a power transistor  301  of the power providing circuit  300 . A drain-source resistance of the power transistor  201 ,  301  is controlled using a gate voltage  325  which is applied to a gate of the power transistor  201 ,  301 . The current is provided to the load  106  at an output voltage  321 . In addition, the method  500  comprises, subject to a drop of the output voltage  321 , coupling  502  the gate of the power transistor  201 ,  301  with a source of the power transistor  201 ,  301 , to put the power transistor  201 ,  301  in an off-state. The coupling  502  may be achieved directly using a short circuit control transistor  403 , thereby bypassing other circuitry (e.g. current limit circuitry) of the power providing circuit  300 . 
     As such, short circuit protection circuitry  400  has been described which allows for an efficient and reliable protection of a power providing circuit  300  subject to a “short circuit” condition, notably in combination with regulated current limit circuitry. 
     It should be noted that the description and drawings merely illustrate the principles of the proposed methods and systems. Those skilled in the art will be able to implement various arrangements that, although not explicitly described or shown herein, embody the principles of the invention and are included within its spirit and scope. Furthermore, all examples and embodiment outlined in the present document are principally intended expressly to be only for explanatory purposes to help the reader in understanding the principles of the proposed methods and systems. Furthermore, all statements herein providing principles, aspects, and embodiments of the invention, as well as specific examples thereof, are intended to encompass equivalents thereof.