Patent Publication Number: US-2012027124-A1

Title: Base station, communication system including base station and transmission method

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     This application is a continuation of U.S. patent application Ser. No. 11/835,906, filed on Aug. 8, 2007, which is a continuation of U.S. patent application Ser. No. 10/704,653, filed Nov. 12, 2003, now U.S. Pat. No. 7,280,840, which issued Oct. 9, 2007, which claims the benefit of Japanese Application Nos. 2002-329453, filed Nov. 13, 2002; 2002-374393, filed Dec. 25, 2002; 2003-018761, filed Jan. 28, 2003 and 2003-366249, filed Oct. 27, 2003, the contents of which are expressly incorporated herein by reference in their entireties. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a receiving apparatus, transmitting apparatus, and reception method, and more particularly to a receiving apparatus, transmitting apparatus, and reception method applied to a radio communication system that uses multiple antennas. 
     2. Description of the Related Art 
     To date, intense research and development has been carried out on radio communication systems that use multiple antennas to allow transmission and reception of a greater amount of data in a limited frequency band. Examples of radio communication systems that use multiple antennas are a MIMO (Multiple-Input Multiple-Output) system in which both the transmitting apparatus and receiving apparatus are equipped with a plurality of antennas, and a MISO (Multiple-Input Single-Output) system in which the transmitting apparatus is equipped with a plurality of antennas and the receiving apparatus is equipped with a single antenna. 
     In a radio communication system that uses this kind of multi-antenna technology, since modulated signals transmitted from a plurality of antennas are multiplexed on a propagation path and received by an antenna at the receiving end, if demodulation processing including signal separation processing cannot be carried out with high precision the receive data error rate characteristics degrade, and as a result, it is not possible to achieve the original aim of increasing the data transmission speed. 
     Possible ways of improving the precision of separation and demodulation of each modulated signal include increasing the pilot symbols inserted in each modulated signal, but when pilot symbols are increased, propagation efficiency degrades proportionally, with the result that it is not in fact possible to achieve the original aim of increasing the data transmission speed. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide a receiving apparatus, transmitting apparatus, and reception method that make it possible to improve the precision of demodulation including separation processing of each modulated signal and improve receive data error rate characteristics in a radio communication system that uses multiple antennas. 
     The present invention estimates a channel fluctuation value on a propagation path of each modulated signal transmitted from a plurality of antennas, finds an eigenvalue of a channel fluctuation matrix formed as an above channel fluctuation value element for relating each antenna received signal to each modulated signal, and using that eigenvalue, performs receiving antenna selection, combining of modulated signals, and weighting processing on a soft decision decoded value, and demodulates each modulated signal. By this means, it is possible to perform demodulation processing based on the effective reception power of a modulated signal (that is, the essential reception power, of the reception power obtained by the receiving apparatus, that can be effectively used when demodulating each modulated signal) thereby enabling the precision of demodulation of each modulated signal to be improved. 
     Also, in a receiving apparatus of the present invention, a further technique is provided whereby an eigenvalue is found by equalizing the power of each element (channel fluctuation value) of the above-mentioned channel fluctuation matrix. This means make it possible to suppress disruption of the relationship between an eigenvalue and effective reception power occurring due to signal amplification processing or analog-digital conversion processing in the radio section, and to find an eigenvalue that reflects effective reception power far more accurately. The processing that equalizes the power of each element of this channel fluctuation matrix also corresponds to finding eigenvalue approximation using only phase of the channel fluctuation matrix, so that an eigenvalue can be found that accurately reflects effective power with a small amount of computation. 
     Furthermore, a transmitting apparatus of the present invention provides independent control for each antenna of the transmission power of the modulated signal transmitted from each antenna based on information such as a channel fluctuation value and received field strength of each modulated signal fed back from the receiving apparatus. By this means, the effective reception power of each modulated signal can be changed more accurately, enabling the precision of demodulation of each modulated signal in the receiving apparatus to be greatly improved. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other objects and features of the present invention will appear more fully hereinafter from a consideration of the following description taken in connection with the accompanying drawings wherein one example is illustrated by way of example, in which: 
         FIG. 1  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 1 of the present invention; 
         FIG. 2  is a block diagram showing a configuration of a reception unit of a transmitting apparatus of Embodiment 1; 
         FIG. 3  is a drawing showing frame configurations of transmit signals transmitted from a transmission unit of a transmitting apparatus; 
         FIG. 4  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 1; 
         FIG. 5  is a block diagram showing a configuration of a transmission unit of a receiving apparatus of Embodiment 1; 
         FIG. 6  is a drawing showing a frame configuration of a transmit signal transmitted from a transmission unit of a transmitting apparatus; 
         FIG. 7  is a drawing illustrating channel fluctuation between antennas of a transmitting apparatus and receiving apparatus; 
         FIG. 8  is a block diagram showing another sample configuration of a transmission unit of a transmitting apparatus; 
         FIG. 9  is a block diagram showing a configuration of the spreading section in  FIG. 8 ; 
         FIG. 10  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 2; 
         FIG. 11  is a drawing showing frame configurations of transmit signals transmitted from the transmission unit in  FIG. 10 ; 
         FIG. 12  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 2; 
         FIG. 13  is a drawing showing a configuration of the inverse Fourier transform section in  FIG. 10 ; 
         FIG. 14  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 3; 
         FIG. 15  is a block diagram showing a configuration of the antenna selection section in  FIG. 14 ; 
         FIG. 16  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 4; 
         FIG. 17  is a block diagram showing a configuration of the signal processing section in  FIG. 16 ; 
         FIG. 18  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 5; 
         FIG. 19  is a block diagram showing a configuration of the signal processing section in  FIG. 18 ; 
         FIG. 20  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 7; 
         FIG. 21  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 7; 
         FIG. 22  is a drawing showing the signal point arrangement in the IQ plane of a BPSK modulated signal; 
         FIG. 23  is a drawing provided for explanation of a BPSK modulated signal soft decision value; 
         FIG. 24  is a block diagram showing another sample configuration of a reception unit of a receiving apparatus of Embodiment 7; 
         FIG. 25  is a drawing provided for explanation of calculation of the distance between a reception point and a candidate point; 
         FIG. 26  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 8; 
         FIG. 27  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 8; 
         FIG. 28  is a block diagram showing another sample configuration of a reception unit of a receiving apparatus of Embodiment 8; 
         FIG. 29  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 9; 
         FIG. 30  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 9; 
         FIG. 31  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 10; 
         FIG. 32  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 10; 
         FIG. 33  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 11; 
         FIG. 34  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 12; 
         FIG. 35  is a drawing showing space-time code frame configurations; 
         FIG. 36  is a drawing showing the relationship between transmitting antennas and a receiving antenna when using space-time coding; 
         FIG. 37  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 13; 
         FIG. 38  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 13; 
         FIG. 39  is a block diagram showing a configuration of the antenna selection section in  FIG. 38 ; 
         FIG. 40  is a drawing showing frame configurations when space-time code is OFDM modulated and transmitted; 
         FIG. 41  is a drawing showing time-frequency coding frame configurations; 
         FIG. 42  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 15; 
         FIG. 43  is a drawing showing a configuration of the signal processing section in  FIG. 42 ; 
         FIG. 44  is a block diagram showing a configuration of a transmission unit of a transmitting apparatus of Embodiment 17; 
         FIG. 45  is a block diagram showing a configuration of a reception unit of a receiving apparatus of Embodiment 17; 
         FIG. 46  is a block diagram showing a configuration of a signal processing section of Embodiment 18; 
         FIG. 47  is a drawing provided for explanation of calculation of Euclidian distance between a reception point and candidate point; 
         FIG. 48  is a block diagram showing a configuration of a signal processing section of Embodiment 19; and 
         FIG. 49  is a drawing showing simulation results when using the configuration of Embodiment 19. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The inventors of the present invention arrived at the present invention by considering that, in a radio communication system that uses multiple antennas, the demodulation precision of each modulated signal can be improved by not simply performing separation and demodulation of each modulated signal, but performing demodulation processing and transmission processing that takes account of the effective reception power of a received modulated signal (that is, the essential reception power, of the reception power obtained by the receiving apparatus, that can be effectively used when demodulating each modulated signal). 
     In the present invention, an eigenvalue of a channel fluctuation matrix is used as an effective reception power index. A channel fluctuation matrix relates each antenna received signal to each modulated signal, with channel fluctuation values as elements. Generally, a receiving apparatus used in multi-antenna communications finds the inverse matrix of the channel fluctuation matrix and separates each modulated signal from the received signal. 
     In the present invention, an eigenvalue is found from a generally used channel fluctuation matrix in this way, and this is used as an effective reception power index, so that it is possible to find the effective reception power with comparatively little computation and comparatively few configuration additions. 
     In the following embodiments, the following kinds of modes of the present invention are chiefly described. 
     In one mode of the present invention, a transmitting apparatus that transmits a plurality of modulated signals from a plurality of antennas performs modification of the transmission power of the transmitted plurality of modulated signals independently for each antenna. Also, transmission power control is performed using received field strength and channel fluctuation estimated by the communicating party. By this means, data transmission quality can be improved. Specifically, it is possible to perform modulated signal transmission power control so that effective reception power is optimized, thereby enabling the demodulation precision of each modulated signal on the receiving side to be improved. 
     In another mode of the present invention, a receiving apparatus that receives a modulated signal transmitted by an above-described transmitting apparatus is equipped with a received field strength estimation section that estimates the received field strength from the received signal, and feeds back estimated received field strength information to the transmitting apparatus. The receiving apparatus is also equipped with a channel fluctuation estimation section that estimates channel fluctuation of each modulated signal from a received signal, and feeds back estimated channel fluctuation information to the transmitting apparatus. By this means, a transmitting apparatus can perform modulated signal transmission power control based on received field strength information and channel fluctuation information so that effective reception power actually becomes optimal on the receiving side. 
     In yet another mode of the present invention, a transmitting apparatus that transmits modulated signals from a plurality of antennas using a multi-antenna system performs modification of the transmission power of the transmitted plurality of modulated signals independently for each antenna and independently for each carrier. Also, the transmitting apparatus performs this transmission power control using per-carrier received field strength and per-carrier channel fluctuation estimated by the communicating party. By this means, it is possible to perform modulated signal transmission power control so that effective reception power becomes optimal independently for each antenna and independently for each carrier. 
     In yet another mode of the present invention, a receiving apparatus that receives a modulated signal transmitted by an above-described multicarrier transmitting apparatus is equipped with a received field strength estimation section that estimates per-carrier received field strength from the received signal, and feeds back estimated per-carrier received field strength information to the multicarrier transmitting apparatus. The receiving apparatus is also equipped with a channel fluctuation estimation section that estimates channel fluctuation for each carrier from a received signal, and feeds back estimated per-carrier channel fluctuation information to the multicarrier transmitting apparatus. By this means, a multicarrier transmitting apparatus can perform modulated signal transmission power control for each carrier based on per-carrier received field strength information and channel fluctuation information so that effective reception power actually becomes optimal on the receiving side. 
     In yet another mode of the present invention, a receiving apparatus that receives a plurality of modulated signals transmitted from a plurality of antennas with a plurality of receiving antennas greater than the plurality of transmitting antennas, creates a plurality of antenna received signal combinations, forms a channel fluctuation matrix for each combination, creates a channel fluctuation matrix eigenvalue for each combination, selects the antenna received signals of the combination whose eigenvalue minimum power is the greatest, and performs demodulation processing. By this means, each modulated signal can be demodulated using the antenna received signal combination with the greatest modulated signal effective reception power, thereby enabling modulated signal demodulation precision to be improved compared with the case where each modulated signal is demodulated using all antenna received signals. 
     In yet another mode of the present invention, a receiving apparatus that receives a plurality of modulated signals transmitted from a plurality of antennas with a plurality of receiving antennas greater than the plurality of transmitting antennas, creates a plurality of antenna received signal combinations, forms a channel fluctuation matrix for each combination, and calculates creates a channel fluctuation matrix eigenvalue for each combination. The receiving apparatus then separates each modulated signal using each combination of antenna received signals and the channel fluctuation matrix corresponding to that combination, and also weights and combines modulated signals separated in each combination using the channel fluctuation matrix eigenvalues used at the time of separation. By this means, it is possible to perform weighting and combining of each modulated signal according to the modulated signal effective reception power, thereby enabling modulated signal demodulation precision to be improved. 
     In yet another mode of the present invention, a receiving apparatus that receives a plurality of modulated signals subjected to error correction coding and transmitted from a plurality of antennas is equipped with a soft decision value calculation section that finds a channel fluctuation matrix eigenvalue, and finds a soft decision value from this eigenvalue and a received quadrature baseband signal. 
     In yet another mode of the present invention, a receiving apparatus that receives a plurality of modulated signals subjected to error correction coding and transmitted from a plurality of antennas is equipped with a soft decision value calculation section that finds an effective reception level from a reception level and a channel fluctuation matrix eigenvalue, and finds a soft decision value from this effective reception level and a received quadrature baseband signal. 
     By performing calculation by weighting a soft decision value with an effective reception level in this way, it is possible to give a soft decision value an appropriate likelihood, and modulated signal demodulation precision can be improved. 
     In yet another mode of the present invention, when demodulation processing is performed using a channel fluctuation matrix eigenvalue, control of the reception level of the received signal received at each antenna is carried out in common for each antenna. By this means, an eigenvalue is found more exactly, so that demodulation processing can be performed based on an eigenvalue that reflects effective reception power much more accurately, thereby enabling the demodulation precision of each modulated signal to be greatly improved. 
     With reference now to the accompanying drawings, embodiments of the present invention will be explained in detail below. 
     Embodiment 1 
     In Embodiment 1, a transmitting apparatus is described that independently modifies the transmission power of a modulated signal transmitted from each antenna. 
       FIG. 1  shows a sample configuration of the transmission unit of a transmitting apparatus according to this embodiment, as provided in a radio base station (hereinafter referred to simply as “base station”), for example. Modulation section  102  of transmission unit  100  has a transmit digital signal  101  and timing signal  122  as input, forms a transmit quadrature baseband signal  103  by executing orthogonal modulation processing such as QPSK (Quadrature Phase Shift Keying) or 16 QAM (Quadrature Amplitude Modulation) on transmit digital signal  101  and performing frame configuration in accordance with timing signal  122  (FIG.  3 (A)), and outputs this transmit quadrature baseband signal  103 . A spreading section  104  has transmit quadrature baseband signal  103  as input, forms a spread signal  105  by executing spreading processing on this transmit quadrature baseband signal  103  using a predetermined spreading code, and outputs this spread signal  105 . A radio section  106  has spread signal  105  as input, forms a modulated signal  107  by executing predetermined radio processing such as digital-analog conversion processing and up-conversion on spread signal  105 , and outputs this modulated signal  107 . 
     A transmission power modification section  108  has modulated signal  107 , a coefficient  125  found from the reception power, and a coefficient  124  found from an eigenvalue as input, obtains a transmit signal  109  by multiplying modulated signal  107  by coefficients  125  and  124 , and outputs this transmit signal  109 . By this means, the transmission power of modulated signal  107  is determined based on the reception power and eigenvalue. Transmit signal  109  is output as a radio wave from an antenna  110 . 
     Modulation section  112  has a transmit digital signal  111  and timing signal  122  as input, forms a transmit quadrature baseband signal  113  by executing orthogonal modulation processing such as QPSK or 16 QAM on transmit digital signal  111  and performing frame configuration in accordance with timing signal  122  (FIG.  3 (B)), and outputs this transmit quadrature baseband signal  113 . A spreading section  114  has transmit quadrature baseband signal  113  as input, forms a spread signal  115  by executing spreading processing on this transmit quadrature baseband signal  113  using a predetermined spreading code, and outputs this spread signal  115 . Spreading section  114  performs spreading processing using a different spreading code from that used by spreading section  104 . A radio section  116  has spread signal  115  as input, forms a modulated signal  117  by executing predetermined radio processing such as digital-analog conversion processing and up-conversion on spread signal  115 , and outputs this modulated signal  117 . 
     A transmission power modification section  118  has modulated signal  117 , a coefficient  126  found from the reception power, and coefficient  124  found from an eigenvalue as input, obtains a transmit signal  119  by multiplying modulated signal  117  by coefficients  125  and  124 , and outputs this transmit signal  119 . By this means, the transmission power of modulated signal  117  is determined based on the reception power and eigenvalue. Transmit signal  119  is output as a radio wave from an antenna  120 . 
     Thus, in transmission unit  100  provided in a transmitting apparatus according to this embodiment, it is possible to modify independently the transmission power of modulated signals transmitted from antennas  110  and  120 . 
       FIG. 2  shows a sample configuration of the reception unit of a transmitting apparatus according to this embodiment. Reception unit  200  is provided in the same base station as transmission unit  100  shown in  FIG. 1 . Radio section  203  of reception unit  200  has a received signal  202  received by an antenna  201  as input, forms a received quadrature baseband signal  204  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  202 , and outputs this received quadrature baseband signal  204 . A demodulation section  205  has received quadrature baseband signal  204  as input, forms a received digital signal  206  by executing orthogonal demodulation processing such as QPSK demodulation or 16 QAM demodulation on received quadrature baseband signal  204 , and outputs this received digital signal  206 . 
     A data separation section  207  has received digital signal  206  as input, separates received digital signal  206  into data  208 , field strength estimation information  209 , and channel fluctuation estimation information  210 , and outputs this data  208 , field strength estimation information  209 , and channel fluctuation estimation information  210 . 
     A reception power based coefficient calculation section  211  has field strength estimation information  209  as input, calculates coefficients  125  and  126  to be used by transmission power modification sections  108  and  118  of transmission unit  100  based on this field strength estimation information  209 , and sends coefficients  125  and  126  to transmission power modification sections  108  and  118 . The method of finding these coefficients  125  and  126  will be described in detail later herein. 
     An eigenvalue based coefficient calculation section  214  has channel fluctuation estimation information  210  as input, calculates coefficient  124  to be used by transmission power modification sections  108  and  118  of transmission unit  100  based on this channel fluctuation estimation information  210 , and sends coefficient  124  to transmission power modification sections  108  and  118 . The method of finding this coefficient  124  will be described in detail later herein. 
       FIG. 3  shows sample frame configurations on the time axis of transmit signals  109  (spread signal A) and  119  (spread signal B) transmitted from antennas  110  and  120  of transmission unit  100 . Spread signal A shown in  FIG. 3(A)  and spread signal B shown in  FIG. 3(B)  are transmitted from antennas  110  and  120  simultaneously. Channel estimation symbols  301  of spread signal A and channel estimation symbols  301  of spread signal B are, for example, mutually orthogonalized codes, and items that can be separated in the reception unit of a terminal are used for this purpose. By this means, a terminal reception unit can estimate channel fluctuation of signals transmitted from antennas  110  and  120  based on channel estimation symbols  301  contained in spread signals A and B. 
       FIG. 4  shows a sample configuration of the reception unit of a receiving apparatus according to this embodiment. Reception unit  400  is provided in a communication terminal, and receives and demodulates a signal transmitted from transmission unit  100  in  FIG. 1 . Radio section  403  of reception unit  400  has a received signal  402  received by an antenna  401  as input, forms a received quadrature baseband signal  404  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  402 , and outputs this received quadrature baseband signal  404 . A despreading section  405  has received quadrature baseband signal  404  as input, forms a despread received quadrature baseband signal  406  by executing de spreading processing using the same spreading code as that used by spreading section  104  and spreading section  114  in  FIG. 1  on received quadrature baseband signal  404 , and outputs this despread received quadrature baseband signal  406 . 
     A spread signal A channel fluctuation estimation section  407  has despread received quadrature baseband signal  406  as input, estimates channel fluctuation of spread signal A (the spread signal transmitted from antenna  110 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  408 . By this means, channel fluctuation between antenna  110  and antenna  401  is estimated. A spread signal B channel fluctuation estimation section  409  has despread received quadrature baseband signal  406  as input, estimates channel fluctuation of spread signal B (the spread signal transmitted from antenna  120 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  410 . By this means, channel fluctuation between antenna  120  and antenna  401  is estimated. 
     Radio section  413  has a received signal  412  received by an antenna  411  as input, forms a received quadrature baseband signal  414  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  412 , and outputs this received quadrature baseband signal  414 . A despreading section  415  has received quadrature baseband signal  414  as input, forms a despread received quadrature baseband signal  416  by executing despreading processing using the same spreading code as that used by spreading section  104  and spreading section  114  in  FIG. 1  on received quadrature baseband signal  414 , and outputs this despread received quadrature baseband signal  416 . 
     A spread signal A channel fluctuation estimation section  417  has despread received quadrature baseband signal  416  as input, estimates channel fluctuation of spread signal A (the spread signal transmitted from antenna  110 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  418 . By this means, channel fluctuation between antenna  110  and antenna  411  is estimated. A spread signal B channel fluctuation estimation section  419  has despread received quadrature baseband signal  416  as input, estimates channel fluctuation of spread signal B (the spread signal transmitted from antenna  120 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  420 . By this means, channel fluctuation between antenna  120  and antenna  411  is estimated. 
     A signal processing section  421  has received quadrature baseband signals  406  and  416 , spread signal A channel fluctuation estimation signals  408  and  418 , and spread signal B channel fluctuation estimation signals  410  and  420  as input, and outputs a spread signal A received quadrature baseband signal  422  and spread signal B received quadrature baseband signal  423  by performing computation using an inverse matrix of a channel fluctuation matrix with channel fluctuation estimation values  408 ,  410 ,  418 , and  420  as elements. Details of this channel fluctuation matrix will be given later herein. 
     A received field strength estimation section  424  has received quadrature baseband signals  406  and  416  as input, finds the received field strength of these signals, and outputs received field strength estimation information  425 . In this embodiment, the received field strength is found from received quadrature baseband signals, but this is not a limitation, and the received field strength may also be found from a received signal. Also, the received field strength may be found separately for spread signal A and spread signal B, or the combined wave received field strength may be found. 
     A channel fluctuation information generation section  426  has spread signal A channel fluctuation estimation signals  408  and  418 , and spread signal B channel fluctuation estimation signals  410  and  420 , as input, and forms and outputs channel fluctuation estimation information  427 . 
       FIG. 5  shows a sample configuration of the transmission unit of a receiving apparatus according to this embodiment. Transmission unit  500  is provided in the same communication terminal as reception unit  400 . information generation section  504  of transmission unit  500  has data  501 , received field strength estimation information  425 , and channel fluctuation estimation information  427  as input, arranges these in a predetermined sequence, and outputs a transmit digital signal  505 . A modulated signal generation section  506  has transmit digital signal  505  as input, forms a modulated signal  507  by executing modulation processing on transmit digital signal  505 , and outputs this modulated signal  507 . A radio section  508  has modulated signal  507  as input, forms a transmit signal  509  by executing predetermined radio processing such as digital-analog conversion processing and up-conversion on modulated signal  507 , and outputs this transmit signal  509 . Transmit signal  509  is output as a radio wave from an antenna  510 . 
       FIG. 6  shows a sample frame configuration of a transmit signal transmitted from transmission unit  500 . In  FIG. 6 , reference numeral  601  denotes channel fluctuation estimation information symbols, reference numeral  602  denotes field strength estimation information symbols, and reference numeral  603  denotes data symbols. 
       FIG. 7  shows an example of the relationship between transmit signals and received signals. Modulated signal Ta(t) transmitted from transmitting antenna  110  is received by antennas  401  and  402  after being subjected to channel fluctuations h 11 ( t ) and h 12 ( t ). Modulated signal Tb(t) transmitted from transmitting antenna  120  is received by antennas  401  and  402  after being subjected to channel fluctuations h 21 ( t ) and h 22 ( t ). 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail using  FIG. 1  through  FIG. 7 . 
     First, the transmission operation of a base station (transmitting apparatus) will be described. An important operation by transmission unit  100  of the base station apparatus shown in  FIG. 1  is to control the transmission power of modulated signals transmitted from antennas  110  and  120  independently at antennas  110  and  120 . For this purpose, transmit signals are multiplied by a coefficient in transmission power modification sections  108  and  118  in transmission unit  100 . 
     The operation of transmission power modification section  108  will be described in detail here . If the value of multiplication coefficient  125  found from the reception power is designated Ca, modulated signal  107  is designated Xa(t), and coefficient  124  found from an eigenvalue is designated D, transmission power modification section  108  controls transmission power Xa′(t) of transmit signal  109  as shown by the following equation. 
       [Equation 1] 
         Xa ′( t )= Ca×D×Xa ( t )  (1)
 
     Similarly, if the value of multiplication coefficient  126  found from the reception power is designated Cb, modulated signal  117  is designated Xb(t), and coefficient  124  found from an eigenvalue is designated D, transmission power modification section  118  controls transmission power Xb′(t) of transmit signal  119  as shown by the following equation. 
       [Equation 2] 
         Xb ′( t )= Cb×D×Xb ( t )  (2)
 
     Performing transmission power control independently for each transmitting antenna in this way enables reception quality to be improved. Also, reception quality can be much more effectively improved by performing multiplication by coefficient  124  value D found from an eigenvalue in common in transmission power modification sections  108  and  118  of both transmitting antennas. This is because a coefficient obtained from an eigenvalue corresponds to the effective received field strength of a receiving terminal (the actual reception field strength, of the reception field strength obtained by a terminal, that can be effectively used). 
     Moreover, reception quality can be much more effectively improved by performing multiplication independently by a coefficient found from reception power in transmission power modification sections  108  and  118  of both transmitting antennas. This is because a coefficient obtained from reception power corresponds to transmission power control for improving the received field strength of each modulated signal at an antenna of a receiving terminal. 
     Next, the reception operation of a base station (transmitting apparatus) will be described. If, as shown in  FIG. 7 , t indicates time, the modulated signal from antenna  110  is designated Ta(t), the modulated signal from antenna  120  is designated Tb(t), the received signal at antenna  401  is designated R 1 ( t ), the received signal at antenna  402  is designated R 2 ( t ), and channel fluctuations are designated h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ), the relationship shown by the following equation applies. That is to say, antenna received signals R 1 ( t ) and R 2 ( t ), and modulated signals Ta(t) and Tb(t), can be related by means of a channel fluctuation matrix with channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) as elements. 
     
       
         
           
             
               
                 
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     Reception power based coefficient calculation section  211  provided in reception unit  200  of the base station (transmitting apparatus) in  FIG. 2  determines coefficients  125  and  126  using field strength estimation information  209  received from the terminal—that is, the received field strengths of R 1 ( t ) and R 2 ( t )—and channel fluctuation estimation information  210 —that is, h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ). 
     For example, coefficient  125  is found from h 11 ( t ) and h 21 ( t ) estimates. This is because h 11 ( t ) and h 12 ( t ) are fluctuation values determined by the transmission power of the signal output from antenna  110  in  FIG. 1 . Similarly, coefficient  126  is found from h 12 ( t ) and h 22 ( t ) estimates, because h 12 ( t ) and h 22 ( t ) are fluctuation values determined by the transmission power of the signal output from antenna  120  in  FIG. 1 . 
     That is to say, the received field strength of R 1 ( t ) and R 2 ( t ) is the field strength of a signal in which both the signal from antenna  110  and the signal from antenna  120  are combined, and therefore if coefficients  125  and  126  are determined based only on that received field strength, this will be insufficient to adjust the signal power from each antenna appropriately. Thus, in this embodiment, in addition to the received field strength, coefficients  125  and  126  for controlling the signal power transmitted from antennas  110  and  120  are determined using channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) at the time of reception of each transmit signal. By this means, the power at the time of reception of each signal transmitted from antennas  110  and  120  can be made appropriate. 
     To given an explanation in concrete terms, when the received field strength is low, the values of coefficients  125  and  126  are naturally made larger so that transmission power increases. Also, the smaller the magnitude of channel fluctuation values h 11 ( t ) and h 21 ( t ), the larger the value of coefficient  125  used by antenna  110  is made. Similarly, the smaller the magnitude of channel fluctuation values h 12 ( t ) and h 22 ( t ), the larger the value of coefficient  126  used by antenna  120  is made. 
     Eigenvalue based coefficient calculation section  214  calculates an eigenvalue of the Equation (3) channel fluctuation matrix with channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) received from the terminal as elements, and finds coefficient  124  based on the value with the lowest power among the eigenvalue power figures. 
     Calculation methods for finding an eigenvalue here include, for example, the Jacobi method, Givens method, Housefolde method, QR method, QL method, QL method with implicit shift, and inverse iteration method, any of which may be used in the present invention. Also, eigenvalue power is a value expressed by a 2 +b 2  when an eigenvalue is expressed in the form a+bj (where a and b are real numbers and j is an imaginary number). The same applies to other embodiments described hereinafter. 
     Next, the reception operation of a communication terminal (receiving apparatus) will be described. Spread signal A channel fluctuation estimation section  407  of reception unit  400  in  FIG. 4  estimates spread signal A channel fluctuation—that is, h 11 ( t ) in Equation (3)—from spread signal A channel estimation symbols  301  shown in  FIG. 3(A) , and outputs the estimation result as spread signal A channel fluctuation estimation signal  408 . Spread signal B channel fluctuation estimation section  409  estimates spread signal B channel fluctuation—that is, h 12 ( t ) in Equation (3)—from spread signal B channel estimation symbols  301  shown in  FIG. 3(B) , and outputs the estimation result as spread signal B channel fluctuation estimation signal  410 . 
     Spread signal A channel fluctuation estimation section  417  estimates spread signal A channel fluctuation—that is, h 21 ( t ) in Equation (3)—from spread signal A channel estimation symbols  301  shown in  FIG. 3(A) , and outputs the estimation result as spread signal A channel fluctuation estimation signal  418 . Spread signal B channel fluctuation estimation section  419  estimates spread signal B channel fluctuation—that is, h 22 ( t ) in Equation (3)—from spread signal B channel estimation symbols  301  shown in  FIG. 3(B) , and outputs the estimation result as spread signal B channel fluctuation estimation signal  420 . 
     Signal processing section  421  finds spread signal A and B received quadrature baseband signals  422  and  423  by performing an inverse matrix operation that multiplies the inverse matrix of the channel fluctuation matrix by both sides in Equation (3). By this means, received quadrature baseband signal  422  and received quadrature baseband signal  423  are separated. Channel fluctuation information generation section  426  has spread signal A channel fluctuation estimation signals  408  and  418 , spread signal B channel fluctuation estimation signals  410  and  420 , and estimated channel fluctuations h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) as input, and outputs these as channel fluctuation estimation information  427 . 
     Thus, according to the above configuration, in a transmitting apparatus that performs multi-antenna transmission it is possible to make the received field strength at the time of reception of each modulated signal appropriate, and thus improve the reception quality of each modulated signal, by receiving from the communicating station channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) at the time of reception of each modulated signal transmitted from antennas  110  and  120 , and independently controlling at antennas  110  and  120  the transmission power of modulated signals transmitted from antennas  110  and  120  based on these channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ). 
     In addition, by controlling transmission power in consideration of an eigenvalue of a channel fluctuation matrix with channel fluctuation values h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) as elements, the effective received field strength can be increased, enabling the reception quality of each modulated signal to be greatly improved. 
     In the above-described embodiment, a case has been described in which coefficients  124 ,  125 , and  126  for controlling the transmission power of antennas  110  and  120  are decided by a base station—that is, on the transmitting side—but the present invention is not limited to this, and it is also possible for coefficients  124 ,  125 , and  126  to be decided by a terminal—that is, on the receiving side—and for the decided coefficients to be fed back to the transmitting side . This also applies to other embodiments described hereinafter. 
     Also, in the above-described embodiment, a case has been described in which the number of antennas is two and the number of multiplexed modulated signals is two, but the present invention is not limited to this, and the present invention can be widely applied to cases where a plurality of antennas are used and a different modulated signal is transmitted from each antenna. It is also possible, for example, for one antenna (for example, antenna  110 ) that transmits a modulated signal to be configured from a plurality of antennas, as with an adaptive array antenna. This also applies to other embodiments described hereinafter. 
     Moreover, in the above-described embodiment, received field strength has been mentioned, but the present invention may also be similarly implemented with reception level, reception strength, reception power, reception amplitude, carrier power to noise power, or the like, substituted for received field strength. This also applies to other embodiments described hereinafter. 
     Furthermore, in the above-described embodiment, symbols transmitted for estimating channel fluctuation are referred to as channel estimation symbols  301  ( FIG. 3 ), but channel estimation symbols  301  may also be referred to as pilot symbols, a preamble, control symbols, known symbols, or a unique word, or may be referred to by another name. Also, channel fluctuation estimation information symbols  601  and field strength estimation information symbols  602  in  FIG. 6  may also be referred to as control symbols, or may be referred to by another name. In other words, the present invention can be implemented in the same way as in the above-described embodiment even if these symbols are used. This also applies to other embodiments described hereinafter. 
     Moreover, in the above-described embodiment, a spread spectrum communication system has been described by way of example, but this is not a limitation, and the present invention can be similarly implemented in a single-carrier system that does not have a spreading section, or an OFDM system, for example. In the case of a single-carrier system, the configuration does not include spreading sections  104  and  114  ( FIG. 1 ) or despreading sections  405  and  415  ( FIG. 4 ). A case in which the present invention is applied to an OFDM system is described in detail in Embodiment 2. 
     Furthermore, the configurations of a transmitting apparatus and receiving apparatus of the present invention are not limited to the configurations in  FIG. 1 ,  FIG. 2 ,  FIG. 4 , and  FIG. 5 . For example, in the above-described embodiment a case has been described in which transmission power modification sections  108  and  118  are provided, and transmission power of modulated signals transmitted from antennas  110  and  120  is controlled independently by these transmission power modification sections at antennas  110  and  120  based on coefficient  124  found from an eigenvalue and coefficients  125  and  126  found from reception power, but it is essential only that the modulated signal of each antenna be controlled independently, and the configuration is not limited to that shown in  FIG. 1 . 
       FIG. 8  shows another sample configuration of the transmission unit of a base station according to this embodiment. In  FIG. 8 , parts that operate in the same way as in transmission unit  100  in  FIG. 1  are assigned the same codes as in  FIG. 1 . The difference between transmission unit  700  in  FIG. 8  and transmission unit  100  in  FIG. 1  is that, whereas transmission unit  100  in  FIG. 1  controls the power of modulated signals transmitted from each antenna by means of transmission power modification sections  108  and  118 , transmission unit  700  in  FIG. 8  controls the power of modulated signals transmitted from each antenna by means of spreading sections  701  and  702 . 
     Specifically, spreading section  701  has transmit quadrature baseband signal  103 , coefficient  125  found from reception power, and coefficient  124  found from an eigenvalue as input, and outputs spread signal  105  of power in accordance with these coefficients  125  and  124 . Similarly, spreading section  702  has transmit quadrature baseband signal  113 , coefficient  126  found from reception power, and coefficient  124  found from an eigenvalue as input, and output s spread signal  115  of power in accordance with these coefficients  126  and  124 . 
       FIG. 9  shows a sample configuration of spreading sections  701  and  702 . A spreading function section  804  has channel X transmit quadrature baseband signal  801 , channel Y transmit quadrature baseband signal  802 , and channel Z transmit quadrature baseband signal  803  as input, forms a channel X spread signal  805 , channel Y spread signal  806 , and channel Z spread signal  807  by performing spreading processing on these signals using different spreading codes, and outputs spread signals  805 ,  806 , and  807 . Here, a channel X signal denotes a signal destined for terminal X, a channel Y signal denotes a signal destined for terminal Y, and a channel Z signal denotes a signal destined for terminal Z. That is to say, transmission unit  700  outputs spread modulated signals destined for three terminals, X, Y, and Z, respectively from antennas  110  and  120 . 
     A coefficient multiplication function section  810  has channel X spread signal  805 , channel Y spread signal  806 , channel Z spread signal  807 , coefficient  125  ( 126 ) found from reception power, and coefficient  124  found from an eigenvalue as input, forms a post-coefficient-multiplication channel X spread signal  811 , post-coefficient-multiplication channel Y spread signal  812 , and post-coefficient-multiplication channel Z spread signal  813  by performing coefficient multiplication in accordance with these coefficients  125  ( 126 ) and  124 , and outputs these signals  811 ,  812 , and  813 . 
     Here, coefficient  125  ( 126 ) found from reception power and coefficient  124  found from an eigenvalue multiplied by channel X spread signal  805  are found based on received field strength estimation information and channel fluctuation estimation information sent from terminal X; coefficient  125  ( 126 ) found from reception power and coefficient  124  found from an eigenvalue multiplied by channel Y spread signal  806  are found based on received field strength estimation information and channel fluctuation estimation information sent from terminal Y; and coefficient  125  ( 126 ) found from reception power and coefficient  124  found from an eigenvalue multiplied by channel Z spread signal  807  are found based on received field strength estimation information and channel fluctuation estimation information sent from terminal Z. 
     An addition function section  814  adds post-coefficient-multiplication channel X spread signal  811 , post-coefficient-multiplication channel Y spread signal  812 , and post-coefficient-multiplication channel Z spread signal  813 , and outputs the result as spread signal  105  ( 115 ). 
     In this way, transmission unit  700  simultaneously generates transmit signals destined for a plurality of terminals. At this time, transmission unit  700  can control transmission power independently for each antenna and independently for the modulated signals destined for each terminal by receiving field strength estimation information and channel fluctuation estimation information from each terminal, finding a coefficient found from reception power and a coefficient found from an eigenvalue for each terminal, and multiplying these coefficients differing for each terminal by the spread modulated signal destined for each terminal. As a result, when modulated signals destined for a plurality of terminals are transmitted from a plurality of antennas, it is possible to optimize the effective reception power at all of the plurality of terminals, and improve the reception quality of all of the plurality of terminals without reducing transmission speed. 
     Thus, according to this embodiment, by receiving information constituting an effective reception power index such as channel fluctuation information and received field strength information from a receiving apparatus as feedback information, and modifying the reception power of the modulated signal transmitted from each antenna independently for each antenna based on this information, it is possible to increase the effective reception power of the modulated signal transmitted from each antenna, and to implement a transmitting apparatus that enables modulated signal reception quality to be improved. 
     Embodiment 2 
     In this embodiment, a transmitting apparatus is described that modifies the transmission power of a modulated signal transmitted from each antenna independently at each antenna and independently for each carrier. 
       FIG. 10  shows a sample configuration of the transmission unit of a transmitting apparatus according to this embodiment. Transmission unit  1000  is provided in a base station apparatus, for example. The base station reception unit is configured as shown in  FIG. 2 , for example, the transmission unit of a terminal that performs communication with the base station is configured as shown in  FIG. 5 , for example, and the frame configuration of a transmit signal transmitted from the terminal transmission unit is as shown in  FIG. 6 , for example. As these have already been described in Embodiment 1, a description thereof is omitted here. 
     In transmission unit  1000 , transmit digital signal  101  and timing signal  122  are input to modulation section  102 , a transmit quadrature baseband signal group  103  is formed by executing orthogonal modulation processing such as QPSK or 16 QAM on transmit digital signal  101  and performing frame configuration in accordance with timing signal  122  (FIG.  11 (A)), and transmit orthogonal baseband group  103  is output. An IDFT  1001  has transmit orthogonal baseband group  103 , coefficient  125  found from reception power, and coefficient  124  found from an eigenvalue as input, modifies the transmission power based on coefficients  125  and  124  and also performs an inverse Fourier transform, and outputs a post-inverse-Fourier-transform signal  1002 . 
     Similarly, in transmission unit  1000 , transmit digital signal  111  and timing signal  122  are input to modulation section  112 , a transmit quadrature baseband signal group  113  is formed by executing orthogonal modulation processing such as QPSK or 16 QAM on transmit digital signal  111  and performing frame configuration in accordance with timing signal  122  (FIG.  11 (B)), and transmit orthogonal baseband group  113  is output. An IDFT  1003  has transmit orthogonal baseband group  113 , coefficient  126  found from reception power, and coefficient  124  found from an eigenvalue as input, modifies the transmission power based on coefficients  126  and  124  and also performs an inverse Fourier transform, and outputs a post-inverse-Fourier-transform signal  1004 . 
       FIG. 11  shows sample frame configurations of modulated signals transmitted from transmission unit  1000 .  FIG. 11(A)  shows the frame configuration of a signal transmitted from antenna  110  (channel A), and  FIG. 11(B)  shows the frame configuration of a signal transmitted from antenna  120  (channel B). In this example, estimation symbols  1101  are transmitted at specific time  1  arranged on all subcarriers, and information symbols  1102  are transmitted at other times  2  through  9 . 
       FIG. 12  shows a sample configuration of the reception unit of a receiving apparatus according to this embodiment. Reception unit  1200  is provided in a communication terminal, and receives and demodulates signals transmitted from transmission unit  1000  in  FIG. 10 . Radio section  1203  of reception unit  1200  has a received signal  1202  received by an antenna  1201  as input, forms a received quadrature baseband signal  1204  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  1202 , and outputs this received quadrature baseband signal  1204 . A Fourier transform section (dft)  1205  has received quadrature baseband signal  1204  as input, forms a post-Fourier-transform signal  1206  by executing Fourier transform processing on received quadrature baseband signal  1204 , and outputs this post-Fourier-transform signal  1206 . 
     A channel A channel fluctuation estimation section  1207  has post-Fourier-transform signal  1206  as input, estimates channel fluctuation of the channel A signal (the OFDM signal transmitted from antenna  110 ) based on the channel A channel estimation symbols, and outputs a channel fluctuation estimation group signal  1208 . By this means, channel fluctuation between antenna  110  and antenna  1201  is estimated. A channel B channel fluctuation estimation section  1209  has post-Fourier-transform signal  1206  as input, estimates channel fluctuation of the channel B signal (the OFDM signal transmitted from antenna  120 ) based on the channel B channel estimation symbols, and outputs a channel fluctuation estimation group signal  1210 . By this means, channel fluctuation between antenna  120  and antenna  1201  is estimated. 
     A radio section  1213  has a received signal  1212  received by an antenna  1211  as input, forms a received quadrature baseband signal  1214  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  1212 , and outputs this received quadrature baseband signal  1214 . A Fourier transform section (dft)  1215  has received quadrature baseband signal  1214  as input, forms a post-Fourier-transform signal  1216  by executing Fourier transform processing on received quadrature baseband signal  1214 , and outputs this post-Fourier-transform signal  1216 . 
     A channel A channel fluctuation estimation section  1217  has post-Fourier-transform signal  1216  as input, estimates channel fluctuation of the channel A signal (the OFDM signal transmitted from antenna  110 ) based on the channel A channel estimation symbols, and outputs a channel fluctuation estimation group signal  1218 . By this means, channel fluctuation between antenna  110  and antenna  1211  is estimated. A channel B channel fluctuation estimation section  1219  has post-Fourier-transform signal  1216  as input, estimates channel fluctuation of the channel B signal (the OFDM signal transmitted from antenna  120 ) based on the channel B channel estimation symbols, and outputs a channel fluctuation estimation group signal  1220 . By this means, channel fluctuation between antenna  120  and antenna  1211  is estimated. 
     A signal processing section  1221  has post-Fourier-transform signals  1206  and  1216 , channel fluctuation estimation group signals  1208  and  1218 , and channel fluctuation estimation group signals  1210  and  1220  as input, and outputs a channel A received quadrature baseband signal group  1222  and channel B received quadrature baseband signal group  1223  by performing computation using an inverse matrix of a channel fluctuation matrix with channel fluctuation estimation values  1208 ,  1218 ,  1210 , and  1220  as elements. 
     A channel A demodulation section  1224  has channel A received quadrature baseband signal group  1222  as input, forms a received digital signal  1225  by executing demodulation processing corresponding to modulation section  102  of transmission unit  1000  ( FIG. 10 ) on that signal, and outputs received digital signal  1225 . A channel B demodulation section  1226  has channel B received quadrature baseband signal group  1223  as input, forms a received digital signal  1227  by executing demodulation processing corresponding to modulation section  112  of transmission unit  1000  on that signal, and outputs received digital signal  1227 . 
     A received field strength estimation section  1228  has post-Fourier-transform signals  1206  and  1216  as input, finds the received field strength of these signals, and outputs received field strength estimation information  1229 . 
     A channel fluctuation estimation section  1230  has channel A channel fluctuation estimation signal groups  1208  and  1218 , and channel B channel fluctuation estimation signal groups  1210  and  1220 , as input, and forms and outputs channel fluctuation estimation information  1231 . 
       FIG. 13  shows a sample configuration of IDFTs  1001  and  1003  provided in transmission unit  1000  in  FIG. 10 . As IDFT  1001  and IDFT  1003  have the same configuration, IDFT  1001  will be described here. 
     IDFT  1001  has a transmission power modification section  1307 . Transmission power modification section  1307  has a carrier  1  transmit quadrature baseband signal  1301 , carrier  2  transmit quadrature baseband signal  1302 , carrier  3  transmit quadrature baseband signal  1303 , carrier  4  transmit quadrature baseband signal  1304 , coefficient  125  found from reception power, and coefficient  124  found from an eigenvalue as input, and by multiplying carrier transmit quadrature baseband signals  1301  through  1304  by coefficients  125  and  124 , obtains post-coefficient-multiplication carrier  1  transmit quadrature baseband signal  1308 , post-coefficient-multiplication carrier  2  transmit quadrature baseband signal  1309 , post-coefficient-multiplication carrier  3  transmit quadrature baseband signal  1310 , and post-coefficient-multiplication carrier  4  transmit quadrature baseband signal  1311 , and outputs these signals. 
     Coefficient  125  found from reception power and coefficient  124  found from an eigenvalue in this embodiment are found for each carrier. Then transmission power modification section  1307  modifies the transmission power on a carrier-by-carrier basis by multiplying the respective corresponding carrier transmit quadrature baseband signals by coefficients  125  and  124 . 
     An inverse Fourier transform section (IDFT section)  1312  has post-coefficient-multiplication carrier  1  transmit quadrature baseband signal  1308 , post-coefficient-multiplication carrier  2  transmit quadrature baseband signal  1309 , post-coefficient-multiplication carrier  3  transmit quadrature baseband signal  1310 , and post-coefficient-multiplication carrier  4  transmit quadrature baseband signal  1311  as input, obtains a post-inverse-Fourier-transform signal  1313  by executing inverse Fourier transform processing on these signals, and outputs post-inverse-Fourier-transform signal  1313 . 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail. To simplify the explanation, the drawings used in Embodiment 1 ( FIG. 2  and  FIG. 6 ) will be used again here. 
     First, the transmission operation of a base station (transmitting apparatus) will be described. Important operations by transmission unit  1000  of the base station apparatus shown in  FIG. 10  are, firstly, to control the transmission power of OFDM signals transmitted from antennas  110  and  120  independently at antennas  110  and  120 , and secondly, to control transmission power on a carrier-by-carrier basis. For this purpose, transmission unit  1000  performs multiplication by coefficients in IDFTs  1001  and  1003  in order to modify the transmission power of transmit quadrature baseband signal groups  103  and  113 . 
     Details of these operations will be described using  FIG. 13 .  FIG. 13  shows the detailed configuration of IDFTs  1001  and  1003  in  FIG. 10 . Transmit orthogonal baseband groups  103  and  113  in  FIG. 10  correspond to carrier  1  transmit quadrature baseband signal  1301 , carrier  2  transmit quadrature baseband signal  1302 , carrier  3  transmit quadrature baseband signal  1303 , and carrier  4  transmit quadrature baseband signal  1304  in  FIG. 13 , and there is an quadrature baseband signal for each subcarrier. 
     Transmission power modification section  1307  modifies transmission power on a carrier-by-carrier basis by multiplying respective corresponding carrier transmit quadrature baseband signals by coefficients  125  and  124 . That is to say, coefficient  125  found from reception power and eigenvalue  126  comprise coefficients for each carrier. The coefficient multiplication method used by transmission power modification section  1307  is basically as described in Embodiment 1, differing only in that coefficient multiplication is performed on a carrier-by-carrier basis. 
     Next, the reception operation of a base station (transmitting apparatus) will be described. In this embodiment, reception unit  200  in  FIG. 2  receives field strength estimation information  209  for each carrier from a communication terminal (receiving apparatus). Then coefficients  125  and  126  for each carrier are found by reception power based coefficient calculation section  211 , and coefficient  124  for each carrier is found by eigenvalue based coefficient calculation section  214 . Thus, coefficients  124 ,  125 , and  126  for each carrier are found based on field strength estimation information  209  and channel fluctuation estimation information  210  for each carrier sent from a communication terminal (receiving apparatus). The coefficient calculation methods used by reception power based coefficient calculation section  211  and eigenvalue based coefficient calculation section  214  are basically as described in Embodiment 1, differing only in that coefficients are calculated on a carrier-by-carrier basis. 
     Next, the reception operation of a communication terminal (receiving apparatus) will be described. Post-Fourier-transform signals  1206  and  1216  output from Fourier transform sections (dft&#39;s)  1205  and  1215  of reception unit  1200  in  FIG. 12  comprise signals for each carrier. 
     Channel A channel fluctuation estimation section  1207  detects estimation symbols  1101  in  FIG. 11(A)  and estimates channel fluctuation on a carrier-by-carrier basis. That is to say, h 11 ( t ) in Equation (3) is estimated for each carrier, and output as channel A channel fluctuation estimation signal group  1208 . Channel B channel fluctuation estimation section  1209  detects estimation symbols  1101  in  FIG. 11(B)  and estimates channel fluctuation on a carrier-by-carrier basis. That is to say, h 12 ( t ) in Equation (3) is estimated for each carrier, and output as channel B channel fluctuation estimation signal group  1210 . 
     Channel A channel fluctuation estimation section  1217  detects estimation symbols  1101  in  FIG. 11(A)  and estimates channel fluctuation on a carrier-by-carrier basis. That is to say, h 21 ( t ) in Equation (3) is estimated for each carrier, and output as channel A channel fluctuation estimation signal group  1218 . Channel B channel fluctuation estimation section  1219  detects estimation symbols  1101  in  FIG. 11(B)  and estimates channel fluctuation on a carrier-by-carrier basis. That is to say, h 22 ( t ) in Equation (3) is estimated for each carrier, and output as channel B channel fluctuation estimation signal group  1219 . 
     Received field strength estimation section  1228  has post-Fourier-transform signals  1206  and  1216  as input, finds the received field strength on a carrier-by-carrier basis, and outputs received field strength estimation signal  1229 . 
     Channel fluctuation estimation section  1230  has channel fluctuation estimation signal groups  1208  and  1218 , and channel fluctuation estimation signal groups  1210  and  1220 , as input, generates channel fluctuation estimation information for each carrier, and outputs this as channel fluctuation estimation information  1231 . 
     Per-carrier received field strength estimation information and per-carrier channel fluctuation estimation information formed in this way is sent to the base station as feedback information by a transmission unit  500  such as shown in  FIG. 5 . Received field strength estimation information  425  in  FIG. 5  corresponds to received field strength estimation information  1229  in  FIG. 12 , and channel fluctuation estimation information  427  in  FIG. 5  corresponds to channel fluctuation estimation information  1231  in  FIG. 12 . 
     Thus, according to this embodiment, when a multicarrier signal is transmitted from a plurality of antennas, by receiving information constituting an effective reception power index such as per-carrier channel fluctuation information and per-carrier received field strength information from a receiving apparatus as feedback information, and modifying the reception power of the multicarrier signal transmitted from each antenna independently for each antenna and independently for each carrier based on this information, it is possible to increase on a carrier-by-carrier basis the effective reception power of the multicarrier signal transmitted from each antenna, and to implement a transmitting apparatus that enables multicarrier signal reception quality to be improved across all carriers. 
     In this embodiment, a case has been described in which transmission power of each carrier is changed by IDFTs  1001  and  1003 , but transmission power need not be modified by IDFTs  1001  and  1003 , but may instead be modified by modulation sections  102  and  112 , or radio sections  106  and  116 , for example. 
     Also, this embodiment has been described taking OFDM as an example, but the present invention can be similarly implemented for a method that combines OFDM processing and spreading processing (such as OFDM-CDMA, for example). 
     Embodiment 3 
     In this embodiment, a transmitting apparatus is described that receives at a plurality of antennas a plurality of modulated signals transmitted from a plurality of antennas, selects a receiving antenna, and performs received signal demodulation using only a received signal from the selected receiving antenna. 
     Specifically, a plurality of antenna received signal combinations are created, a channel fluctuation matrix is created for each combination, channel fluctuation matrix eigenvalues are calculated for each combination, and antenna received signals of the combination for which the eigenvalue minimum power is greatest are selected, and undergo demodulation processing. 
       FIG. 14  shows a sample configuration of the reception unit of a receiving apparatus according to this embodiment. Parts in  FIG. 14  corresponding to those in  FIG. 4  are assigned the same codes as in  FIG. 4  and detailed descriptions of these parts are omitted. Reception unit  1400  is provided in a communication terminal, for example. Here, it is assumed that the transmission unit of a base station that performs communication with a communication terminal equipped with reception unit  1400  is configured as shown in  FIG. 1 , for example, and signals transmitted from the base station are configured as shown in  FIG. 3 . 
     Reception unit  1400  has three antennas  401 ,  411 , and  1401 , and two modulated signals (spread signal A and spread signal B) transmitted from transmission unit  100  are received by each of antennas  401 ,  411 , and  1401 . 
     Radio section  1403  of reception unit  1400  has a received signal  1402  received by antenna  1401  as input, forms a received quadrature baseband signal  1404  by executing predetermined radio processing such as down-conversion and analog-digital conversion on received signal  1402 , and outputs this received quadrature baseband signal  1404 . A despreading section  1405  has received quadrature baseband signal  1404  as input, forms a despread received quadrature baseband signal  1406  by executing despreading processing using the same spreading code as that used by spreading section  104  and spreading section  114  in  FIG. 1  on received quadrature baseband signal  1404 , and outputs this despread received quadrature baseband signal  1406 . 
     A spread signal A channel fluctuation estimation section  1407  has despread received quadrature baseband signal  1406  as input, estimates channel fluctuation of spread signal A (the spread signal transmitted from antenna  110 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  1408 . By this means, channel fluctuation between antenna  110  and antenna  1401  is estimated. A spread signal B channel fluctuation estimation section  1409  has despread received quadrature baseband signal  1406  as input, estimates channel fluctuation of spread signal B (the spread signal transmitted from antenna  120 ) based on the channel estimation symbols, and outputs a channel fluctuation estimation signal  1410 . By this means, channel fluctuation between antenna  120  and antenna  1401  is estimated. 
     An antenna selection section  1411  has channel A channel fluctuation estimation signals  408 ,  418 , and  1408 , channel B channel fluctuation estimation signals  410 ,  420 , and  1410 , and despread received quadrature baseband signals  406 ,  416 , and  1406  as input, and selects from among these the optimal antenna received signal combination for demodulation. The selection method will be described later herein. Antenna selection section  1411  outputs selected spread signal A channel fluctuation estimation signals  1412  and  1415 , selected spread signal B channel fluctuation estimation signals  1413  and  1416 , and selected despread received quadrature baseband signals  1414  and  1417 . 
       FIG. 15  shows a sample configuration of antenna selection section  1411 . Antenna selection section  1411  has an eigenvalue calculation section  1501  and a signal selection section  1503 . Eigenvalue calculation section  1501  has channel A channel fluctuation estimation signals  408 ,  418 , and  1408 , and channel B channel fluctuation estimation signals  410 ,  420 , and  1410 , as input. That is to say, since three antennas are provided in this embodiment, three sets of channel fluctuation values are input. Then combinations of two sets of the three sets of channel fluctuation values are created (in this embodiment, three combinations), a channel fluctuation matrix is created for each of those combinations, and eigenvalues of each channel fluctuation matrix are calculated. Two sets of signals for inverse matrix calculation are then selected based on the eigenvalue calculation results, and a control signal  1502  indicating which two sets have been selected is output. 
     Signal selection section  1503  has channel A channel fluctuation estimation signals  408 ,  418 , and  1408 , channel B channel fluctuation estimation signals  410 ,  420 , and  1410 , despread received quadrature baseband signals  406 ,  416 , and  1406 , and control signal  1502  as input, and outputs selected spread signal A channel fluctuation estimation signals  1412  and  1415 , selected spread signal B channel fluctuation estimation signals  1413  and  1416 , and selected despread received quadrature baseband signals  1414  and  1417 . 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail. 
     The operation of a base station (transmitting apparatus) is the same as that described in Embodiment 1, transmitting transmit signals in accordance with the frame configurations shown in  FIG. 3 . 
     A communication terminal (receiving apparatus) receives transmit signals at three antennas provided on reception unit  1400  in  FIG. 14 . A special feature here is that the number of antennas is made larger than the number of channels transmitted by the transmitting apparatus, and antenna selection is performed. That is to say, antenna selection section  1411  selects two signal groups from signal groups  406 ,  408 , and  410  obtained by antenna  401 , signal groups  416 ,  418 , and  420  obtained by antenna  411 , and signal groups  1406 ,  1408 , and  1410  obtained by antenna  1401 , and performs separation and demodulation using only the selected signal groups. 
     The signal group selection method at this time will now be described. First, eigenvalue calculation section  1501  shown in  FIG. 15  creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  408 ,  410 ,  418 , and  420  in the relationship in  FIG. 7 , and finds value P 1  with the smallest power among those eigenvalues. Eigenvalue calculation section  1501  also creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  408 ,  410 ,  1408 , and  1410  in the relationship in  FIG. 7  , and finds value P 2  with the smallest power among those eigenvalues. Eigenvalue calculation section  1501  further creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  418 ,  420 ,  1408 , and  1410  in the relationship in  FIG. 7 , and finds value P 3  with the smallest power among those eigenvalues. 
     Eigenvalue calculation section  1501  then searches for the largest value among P 1 , P 2 , and P 3 . If P 1  is the largest, eigenvalue calculation section  1501  outputs a control signal  1502  indicating that signals  408 ,  410 ,  406 ,  418 ,  420 , and  416  are to be selected. That is to say, eigenvalue calculation section  1501  instructs signal selection section  1503  to select the signal groups obtained from antennas  401  and  411  in  FIG. 14 . 
     At this time, signal selection section  1503  outputs signal  408  as signal  1412 , signal  410  as signal  1413 , signal  406  as signal  1414 , signal  418  as signal  1415 , signal  420  as signal  1416 , and signal  416  as signal  1417 . Similarly, if P 2  is the largest the signal groups obtained from antennas  401  and  1401  are selected, and if P 3  is the largest the signal groups obtained from antennas  411  and  1401  are selected. 
     Signal processing section  421  in  FIG. 14  sets up Equation (3) in the relationship in  FIG. 7  using input signals  1412 ,  1413 ,  1414 ,  1415 ,  1416 , and  1417 , and by performing the inverse matrix operation of that equation, separates the signals of each channel and outputs separated channel signals  422  and  423 . 
     By switching receiving antennas based on the channel fluctuation matrix eigenvalue for which power is smallest in this way, it is possible to select the antenna with the best reception quality. By this means, the error rate characteristics of demodulated data can be improved. 
     Eigenvalue minimum power corresponds to the effective reception power of a modulated signal contained in an antenna received signal used to obtain that eigenvalue, and therefore selecting an antenna received signal for which eigenvalue minimum power is greatest is equivalent to selecting an antenna received signal combination for which modulated signal effective reception power is greatest. It is therefore possible to demodulate each modulated signal using a combination of antenna received signals for which modulated signal effective reception power is greatest, enabling modulated signal demodulation precision to be greatly improved compared with the case where each modulated signal is demodulated using all antenna received signals. 
     Thus, according to this embodiment, by creating a plurality of antenna received signal combinations, creating a channel fluctuation matrix for each combination, calculating channel fluctuation matrix eigenvalues for each combination, selecting antenna received signals of the combination for which the eigenvalue minimum power is greatest, and performing demodulation processing, it is possible to implement a receiving apparatus that enables the error rate characteristics of a received plurality of channel signals to be improved. 
     In this embodiment a case has been described in which modulated signals of two channels transmitted from two antennas are received by three antennas, but the number of transmitting antennas and number of receiving antennas are not limited to these numbers. The present invention can be widely applied to cases where a plurality of transmitting antennas are provided, a greater number of receiving antennas are provided, and receiving antennas equal to the number of channels are selected from the plurality of receiving antenna signals. 
     Also, in the above-described embodiment, a spread spectrum communication system has been described by way of example, but this is not a limitation, and the present invention can be similarly implemented in a single-carrier system that does not have a spreading section, or an OFDM system, for example. A case in which the present invention is applied to an OFDM system is described in detail in Embodiment 4. 
     Embodiment 4 
     In this embodiment, a case is described in which the processing described in Embodiment 3 is applied to OFDM communications. A special feature of this embodiment is that the following processing is performed for each subcarrier: a plurality of antenna received signal combinations are created, a channel fluctuation matrix is created for each combination, channel fluctuation matrix eigenvalues are calculated for each combination, and antenna received signals of the combination for which the eigenvalue minimum power is greatest are selected, and undergo demodulation processing. 
       FIG. 16  shows a sample configuration of the reception unit of a receiving apparatus according to this embodiment. Reception unit  1600  of this embodiment has many parts combining Embodiment 2 and Embodiment 3, and therefore parts corresponding to parts in  FIG. 12  described in Embodiment 2 are assigned the same codes as in  FIG. 12 , parts corresponding to parts in  FIG. 14  described in Embodiment 3 are assigned the same codes as in  FIG. 14 , and descriptions of these parts are omitted. 
     Reception unit  1600  is provided in a communication terminal, for example. Here, it is assumed that the transmission unit of a base station that performs communication with a communication terminal equipped with reception unit  1600  is configured as shown in  FIG. 10 , for example, and signals transmitted from the base station are configured as shown in  FIG. 11 . 
     Reception unit  1600  has three antennas  401 ,  411 , and  1401 , and two OFDM signals transmitted from transmission unit  1000  are received by each of antennas  401 ,  411 , and  1401 . A special feature of reception unit  1600  here is that the number of antennas (in this embodiment three) is greater than the number of channels of signals transmitted by transmission unit  1000  (in this embodiment, two). 
     Received signals  402 ,  412 , and  1402  of antennas  401 ,  411 , and  1401  become received quadrature baseband signals  404 ,  414 , and  1404  by undergoing predetermined radio processing such as down-conversion and analog-digital conversion by radio sections  403 ,  413 , and  1403 , respectively. Received quadrature baseband signals  404 ,  414 , and  1404  become post-Fourier-transform signals  1206 ,  1216 , and  1602  by undergoing Fourier transform processing by Fourier transform sections (dft&#39;s)  1205 ,  1215 , and  1601 , respectively. 
     Post-Fourier-transform signals  1206 ,  1216 , and  1602  obtained for each antenna are sent to channel A channel fluctuation estimation sections  1207 ,  1217 , and  1603 , and channel B channel fluctuation estimation sections  1209 ,  1219 , and  1605 , provided for each antenna. Channel A channel fluctuation estimation sections  1207 ,  1217 , and  1603  obtain per-carrier channel fluctuation estimation signal groups  1208 ,  1218 , and  1604  for channel A, and send these to a signal processing section  1607 . 
     Signal processing section  1607  performs processing combining antenna selection section  1411  and signal processing section  421  in  FIG. 14 . That is to say, signal processing section  1607  performs antenna signal selection based on eigenvalue power, and also performs channel signal separation processing using the selected antenna signals. However, signal processing section  1607  of this embodiment differs from reception unit  1400  in  FIG. 14  in that the above antenna signal selection processing and channel signal separation processing are performed on a carrier-by-carrier basis. Signal processing section  1607  has channel A channel fluctuation estimation signal groups  1208 ,  1218 , and  1604 , channel B channel fluctuation estimation signal groups  1210 ,  1220 , and  1606 , and post-Fourier-transform signals  1206 ,  1216 , and  1602  as input, and outputs a channel A received quadrature baseband signal  1608  and channel B received quadrature baseband signal  1609  on which selection processing and separation processing have been executed on a carrier-by-carrier basis. 
       FIG. 17  shows the detailed configuration of signal processing section  1607 . The signal processing section configuration shown in  FIG. 17  is the configuration for performing processing for one carrier, and signal processing section  1607  in  FIG. 16  is actually provided with a circuit as shown in  FIG. 17  for each carrier. 
     Eigenvalue calculation section  1701  has the same function as eigenvalue calculation section  1501  in  FIG. 15  described in Embodiment 3. That is to say, eigenvalue calculation section  1701  creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  1208 - 1 ,  1210 - 1 ,  1218 - 1 , and  1220 - 1  for carrier  1  in  FIG. 11  from among channel fluctuation estimation signal groups  1208 ,  1210 ,  1218 , and  1220 , and finds value P 1  with the smallest power among those eigenvalues. Eigenvalue calculation section  1701  also creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  1208 - 1 ,  1210 - 1 ,  1604 - 1 , and  1606 - 1  for carrier  1  from among channel fluctuation estimation signal groups  1208 ,  1210 ,  1604 , and  1606 , and finds value P 2  with the smallest power among those eigenvalues. Eigenvalue calculation section  1701  further creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  1218 - 1 ,  1220 - 1 ,  1604 - 1 , and  1606 - 1  for carrier  1  from among channel fluctuation estimation signal groups  1218 ,  1220 ,  1604 , and  1606 , and finds value P 3  with the smallest power among those eigenvalues. 
     Eigenvalue calculation section  1701  then searches for the largest value among P 1 , P 2 , and P 3 . If P 1  is the largest, eigenvalue calculation section  1701  outputs a control signal  1702  indicating that signals  1208 - 1 ,  1210 - 1 ,  1206 - 1 ,  1218 - 1 ,  1220 - 1 , and  1216 - 1  are to be selected. That is to say, eigenvalue calculation section  1701  instructs signal selection section  1703  to select the signal groups obtained from antennas  401  and  411  in  FIG. 16 . 
     At this time, signal selection section  1703  outputs signal  1208 - 1  as signal  1704 , signal  1210 - 1  as signal  1705 , signal  1206 - 1  as signal  1706 , signal  1218 - 1  as signal  1707 , signal  1220 - 1  as signal  1708 , and signal  1216 - 1  as signal  1709 . Similarly, if P 2  is the largest the signal groups obtained from antennas  401  and  1401  are selected, and if P 3  is the largest the signal groups obtained from antennas  411  and  1401  are selected. 
     A computation section  1710  sets up Equation (3) in the relationship in  FIG. 7  using input signals  1704  through  1709 , and by performing the inverse matrix operation of that equation, separates the signals of each channel and outputs separated channel A carrier  1  quadrature baseband signal  1608 - 1  and channel B carrier  1  quadrature baseband signal  1609 - 1 . 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail. 
     The operation of a base station (transmitting apparatus) is the same as that described in Embodiment 2, transmitting transmit signals in accordance with the frame configurations shown in  FIG. 11 . 
     A communication terminal (receiving apparatus) receives two channels of OFDM signals at three antennas provided on reception unit  1600  in  FIG. 16 . Reception unit  1600  then estimates channel fluctuation on a channel-by-channel basis and on a carrier-by-carrier basis for reception at each antenna. 
     Reception unit  1600  then performs the following processing for each carrier: creation of a plurality of antenna received signal combinations, creation of a channel fluctuation matrix for each combination, channel fluctuation matrix eigenvalue calculation for each combination, and selection of antenna received signals of the combination for which the eigenvalue minimum power is greatest. In this embodiment, as the number of received OFDM signal channels is two and the number of receiving antennas is three, three combinations are created, and one combination is selected from among these three combinations. 
     Next, reception unit  1600  separates the signals of each channel multiplexed on the propagation path by performing an inverse matrix operation using the selected combination of antenna received signals (channel fluctuation estimation and quadrature baseband signals). Then, lastly, receive data is obtained by demodulating the separated channel signals. 
     As reception unit  1600  selects an antenna received signal for which channel fluctuation matrix eigenvalue minimum power is greatest, separates modulated signals (that is, signals transmitted from different antennas) multiplexed on the propagation path using the selected antenna received signal, and performs demodulation processing in this way on a carrier-by-carrier basis, it is possible to perform signal separation and demodulation processing using the antenna received signal with the greatest effective reception power. 
     With OFDM signals in particular, effective reception power differs greatly from carrier to carrier due to the effects of frequency selective fading, etc. In this embodiment this is taken into consideration, and the optimal antenna received signal combination is selected on a carrier-by-carrier basis by performing antenna selection based on eigenvalues for each carrier. By this means, error rate characteristics can be improved across all carriers. 
     Thus, according to this embodiment, by performing, on a carrier-by-carrier basis, creation of a plurality of antenna received signal combinations, creation of a channel fluctuation matrix for each combination, channel fluctuation matrix eigenvalue calculation for each combination, selection of antenna received signals of the combination for which the eigenvalue minimum power is greatest, and demodulation processing, it is possible to implement a receiving apparatus that enables the error rate characteristics of received OFDM signals of a plurality of channels to be improved across all carriers. 
     In this embodiment a case has been described in which OFDM signals of two channels transmitted from two antennas are received by three antennas, but the number of transmitting antennas and number of receiving antennas are not limited to these numbers. The present invention can be widely applied to cases where a plurality of transmitting antennas are provided, a greater number of receiving antennas are provided, and receiving antennas equal to the number of channels are selected from the plurality of receiving antenna signals. 
     Also, in this embodiment, an OFDM system has been described by way of example, but the present invention can be similarly implemented in a system combining an spread spectrum system as described in Embodiment 3 and an OFDM system, and can also be similarly implemented in a multicarrier system other than OFDM. 
     Embodiment 5 
     In this embodiment, a receiving apparatus is described that receives at a plurality of antennas a plurality of modulated signals transmitted from a plurality of antennas, and performs weighting and combining of received signals received at each receiving antenna based on channel fluctuation matrix eigenvalues. 
     To be specific, firstly, a plurality of antenna received signal combinations are created, a channel fluctuation matrix is created for each combination, and channel fluctuation matrix eigenvalues are calculated for each combination. Then, modulated signals are separated using the antenna received signals of each combination and the channel fluctuation matrix corresponding to that combination, and modulated signals separated in each combination are weighted and combined using the channel fluctuation estimation matrix eigenvalues used at the time of separation. 
       FIG. 18  shows a sample configuration of the reception unit of a receiving apparatus according to this embodiment. Parts in  FIG. 18  corresponding to parts in  FIG. 14  are assigned the same codes as in  FIG. 14 , and descriptions of these parts are omitted. Reception unit  1800  is provided in a communication terminal, for example. Here, it is assumed that the transmission unit of a base station that performs communication with a communication terminal equipped with reception unit  1800  is configured as shown in  FIG. 1 , for example, and signals transmitted from the base station are configured as shown in  FIG. 3 . 
     The difference between reception unit  1400  in  FIG. 14  described in Embodiment 3 and reception unit  1800  of this embodiment is that, whereas reception unit  1400  selects an antenna signal using separation and demodulation based on channel fluctuation matrix eigenvalues, reception unit  1800  of this embodiment weights and combines antenna received signals based on channel fluctuation matrix eigenvalues. Therefore, reception unit  1800  has a signal processing section  1801  instead of antenna selection section  1411  and signal processing section  421  of reception unit  1400 , and performs weighting and combining processing on antenna received signals based on channel fluctuation matrix eigenvalues by means of signal processing section  1801 . 
     That is to say, signal processing section  1801  has three sets of antenna signals—spread signal A channel fluctuation estimation signals  408 ,  418 , and  1408 , spread signal B channel fluctuation estimation signals  410 ,  420 , and  1410 , and despread received quadrature baseband signals  406 ,  416 , and  1406 —as input, creates combinations each of two sets of signals in the same way as in Embodiment 3, creates a channel fluctuation matrix for each combination, and calculates the eigenvalue thereof for each combination. Signal processing section  1801  also separates channel A and channel B signals for each combination by performing channel fluctuation matrix inverse matrix computations for each combination. The channel signals separated on a combination-by-combination basis then undergo weighting and combining using the eigenvalues corresponding to each combination. Signal processing section  1801  then outputs weighted and combined channel signals  422  and  423 . 
       FIG. 19  shows a sample configuration of signal processing section  1801 . Signal processing section  1801  has an eigenvalue calculation section  1901  and a separation/combination section  1903 . Eigenvalue calculation section  1901  has spread signal A channel fluctuation estimation signals  408 ,  418 , and  1408 , and spread signal B channel fluctuation estimation signals  410 ,  420 , and  1410 , as input. That is to say, since three antennas are provided in this embodiment, three sets of channel fluctuation values are input. Then combinations of two sets of the three sets of channel fluctuation values are created (in this embodiment, three combinations), a channel fluctuation matrix is created for each of those combinations, and eigenvalues of each channel fluctuation matrix are calculated. Eigenvalues for each combination are then output as an eigenvalue estimation signal  1902 . 
     Separation/combination section  1903  has spread signal A channel fluctuation estimation signals  408 ,  418 , and  1408 , spread signal B channel fluctuation estimation signals  410 ,  420 , and  1410 , despread received quadrature baseband signals  406 ,  416 , and  1406 , and eigenvalue estimation signal  1902  as input, performs channel signal separation processing on a combination-by-combination basis, and also performs weighting and combining processing on the antenna received signals using eigenvalue estimation signal  1902 . By this means, separation/combination. section  1903  obtains spread signal A received quadrature baseband signal  422  and spread signal B received quadrature baseband signal  423 , which it outputs. 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail. 
     The operation of a base station (transmitting apparatus) is the same as that described in Embodiment 1, transmitting transmit signals in accordance with the frame configurations shown in  FIG. 3 . 
     A communication terminal (receiving apparatus) receives transmit signals at three antennas provided on reception unit  1800  in  FIG. 18 . Reception unit  1800  then estimates channel fluctuation on a channel-by-channel basis for reception at each antenna by means of channel fluctuation estimation sections  407 ,  409 ,  417 ,  419 ,  1407 , and  1409 . 
     Next, reception unit  1800  creates a plurality of antenna received signal combinations, forms a channel fluctuation matrix for each combination, and calculates channel fluctuation matrix eigenvalues for each combination. Reception unit  1800  performs this per-combination eigenvalue calculation processing by means of eigenvalue calculation section  1901 . 
     Specifically, eigenvalue calculation section  1901  creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  408 ,  410 ,  418 , and  420  in the relationship in  FIG. 7 , and finds value P 1  with the smallest power among those eigenvalues. Eigenvalue calculation section  1901  also creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  408 ,  410 ,  1408 , and  1410  in the relationship in  FIG. 7 , and finds value P 2  with the smallest power among those eigenvalues. Eigenvalue calculation section  1901  further creates a channel fluctuation matrix as shown in Equation (3) using channel fluctuation estimation signals  418 ,  420 ,  1408 , and  1410  in the relationship in  FIG. 7 , and finds value P 3  with the smallest power among those eigenvalues. Then eigenvalue calculation section  1901  sends obtained values P 1 , P 2 , and P 3  to separation/combination section  1903  as an eigenvalue estimation signal  1902 . 
     Separation/combination section  1903  first performs channel signal separation processing for each antenna received signal combination. In this embodiment, separation processing is performed for three sets of antenna received signals. That is to say, for the first set, separation/combination. section  1903  sets up Equation (3) in the relationship in  FIG. 7  using input signals  408 ,  410 ,  406 ,  418 ,  420 , and  416 , and performs the inverse matrix operation of that equation. The spread signal A received quadrature baseband signal and spread signal B received quadrature baseband signal thus obtained are designated Ra 1  and Rb 1  respectively. For the second set, separation/combination section  1903  sets up Equation (3) in the relationship in  FIG. 7  using input signals  408 ,  410 ,  406 ,  1408 ,  1410 , and  1406 , and performs the inverse matrix operation of that equation. The spread signal A received quadrature baseband signal and spread signal B received quadrature baseband signal thus obtained are designated Ra 2  and Rb 2  respectively. For the third set, separation/combination section  1903  sets up Equation (3) in the relationship in  FIG. 7  using input signals  418 ,  420 ,  416 ,  1408 ,  1410 , and  1406 , and performs the inverse matrix operation of that equation. The spread signal A received quadrature baseband signal and spread signal B received quadrature baseband signal thus obtained are designated Ra 3  and Rb 3  respectively. 
     Separation/combination section  1903  performs the weighting and combining operations of the following equations using the thus obtained sets of received quadrature baseband signals Ra 1 , Rb 1 , Ra 2 , Rb 2 , Ra 3 , and Rb 3 , and eigenvalues P 1 , P 2 , and P 3  corresponding to each set, thereby obtaining weighted and combined spread signal A received quadrature baseband signal Ra ( 422 ) and spread signal B received quadrature baseband signal Rb ( 423 ). 
     
       
         
           
             
               
                 
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     By thus performing weighting to find the received quadrature baseband signal of each channel, more precise spread signal A and B received quadrature baseband signals  422  and  423  are obtained. This is because eigenvalue power is a value corresponding to effective reception power. Thus, in reception processing of this embodiment, effective use is made of reception levels using channel fluctuation matrix eigenvalue power—that is to say, effective reception levels are found and signal combination is performed based on these effective reception levels. 
     Spread signal A and B received quadrature baseband signals  422  and  423  output from separation/combination section  1903  each undergo orthogonal demodulation processing by a demodulation section (not shown) to become receive data. As a result, it is possible to obtain receive data of each channel with good error rate characteristics. 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas a plurality of modulated signals transmitted from a plurality of antennas, by weighting and combining received signals obtained at each receiving antenna based on channel fluctuation matrix eigenvalues it is possible to weight more heavily an antenna received signal with greater effective reception power, enabling the error rate characteristics of a received plurality of channel signals to be improved. 
     In this embodiment a case has been described in which modulated signals of two channels transmitted from two antennas are received by three antennas, but the number of transmitting antennas and number of receiving antennas are not limited to these numbers. The present invention can be widely applied to cases where a plurality of transmitting antennas are provided, a greater number of receiving antennas are provided, and receiving antennas equal to the number of channels are selected from the plurality of receiving antenna signals. 
     Also, in this embodiment a method has been described whereby channel fluctuation matrix eigenvalue power is taken as a weighting coefficient, and received quadrature baseband signal weighting and combining is performed based on this coefficient, but the present invention is not limited to this. 
     The method according to the present embodiment can be applied to cases where received signals are applied error correction codes such as convolutional code, turbo code, and low density parity code. The decoding in this case is executed by finding a branch metric and a path metric sequentially based on weighted results. 
     For example, channel fluctuation matrix eigenvalue power described in this embodiment may also be used as a weighting coefficient for MLD (Maximum Likelihood Detection) shown in “A simple transmit diversity technique for wireless communications” IEEE Journal on Select Areas in Communications, vol. 16, no. 8, October 1998. Use of channel fluctuation matrix eigenvalue power as a weighting coefficient in demodulation and decoding when performing MLD improves reception quality. A weighting method using an eigenvalue for MLD is described in detail in Embodiment 7 onward. 
     Embodiment 6 
     In this embodiment, a case is described in which the processing described in Embodiment 5 is applied to OFDM communications. A special feature of this embodiment is that the processing described in Embodiment 5 whereby received signals obtained at each receiving antenna are weighted and combined based on channel fluctuation matrix eigenvalues is performed for each carrier. 
       FIG. 18  and  FIG. 19  used in Embodiment 5 will also be used in describing this embodiment. The reception unit of this embodiment has a configuration in which despreading sections  405 ,  415 , and  1405  in  FIG. 18  are replaced by Fourier transform sections (dft&#39;s), channel fluctuation estimation sections  407 ,  409 ,  417 ,  419 ,  1407 , and  1409  in  FIG. 18  are configured so as to estimate signal channel fluctuation on a carrier-by-carrier basis, and signal processing section  1801  in  FIG. 18  is configured so as to weight and combine antenna received signals of each carrier using per-carrier channel fluctuation matrix eigenvalues as weight coefficients. 
     Actually, the kind of configuration shown in  FIG. 19  is provided for each carrier as a signal processing section, and the channel fluctuation matrix eigenvalue based weighting and combining described in Embodiment 5 is performed for each carrier. As a result, the signal error rate characteristics can be improved for all carriers. 
     As also described in Embodiment 4, with OFDM signals, effective reception power differs greatly from carrier to carrier due to the effects of frequency selective fading, etc. In this embodiment this is taken into consideration, and the weight coefficient used in combining is changed on a carrier-by-carrier basis by performing signal combining with eigenvalue power as a weight coefficient on a carrier-by-carrier basis. By this means, error rate characteristics can be improved across all carriers. 
     Thus, according to this embodiment, when OFDM signals transmitted from a plurality of antennas are received at a plurality of antennas and demodulated, by performing processing whereby received signals obtained at each receiving antenna are weighted and combined based on channel fluctuation matrix eigenvalues, as described in Embodiment 5, for each carrier, it is possible to implement a receiving apparatus that enables the error rate characteristics of received OFDM signals of a plurality of channels to be improved across all carriers. 
     In this embodiment a method has been described whereby received quadrature baseband signal weighting and combining is performed on a carrier-by-carrier basis using channel fluctuation matrix eigenvalue power as a weighting coefficient, but the present invention is not limited to this. 
     For example, channel fluctuation matrix eigenvalue power described in this embodiment may also be used as a weighting coefficient for MLD (Maximum Likelihood Detection) shown in “A simple transmit diversity technique for wireless communications” IEEE Journal on Select Areas in Communications, vol. 16, no. 8, October 1998. Use of per-carrier channel fluctuation matrix eigenvalue power as a per-carrier weighting coefficient in demodulation and decoding improves reception quality. MLD is described in detail in Embodiment 9 and Embodiment 10. 
     Embodiment 7 
     In this embodiment, a receiving apparatus is described that receives at a plurality of antennas a plurality of modulated signals transmitted from a plurality of antennas, and performs weighting processing on received signals and demodulates received signals using channel fluctuation matrix eigenvalues and the received field strength of each antenna received signal. 
     Specifically, a soft decision value of each modulated signal after separation is weighted using a channel fluctuation matrix eigenvalue. By this means, a soft decision value can be given an appropriate likelihood according to the effective reception power of the modulated signal. As a result, the error rate characteristics of a received digital signal obtained by a decoding section is improved. 
     First, the configuration of a transmitting apparatus will be described.  FIG. 20  shows an example of the configuration of the transmission unit of a transmitting apparatus according to this embodiment. The difference between transmission unit  2000  of this embodiment and transmission unit  100  in  FIG. 1  is that transmission unit  2000  has error correction coding sections  2001  and  2002 . The rest of the configuration is the same as that of transmission unit  100  in  FIG. 1 , and therefore a description thereof is omitted here. 
     Error correction coding sections  2001  and  2002  have transmit digital signals  101  and  111  as input respectively, obtain error correction coded signals  2003  and  2004  by executing error correction coding processing on transmit digital signals  101  and  111  using convolutional code, and output these signals  2003  and  2004 . 
     Modulation sections  102  and  112  have error correction coded signals  2003  and  2004  as input respectively, and executed modulation processing on error correction coded signals  2003  and  2004 . In this embodiment, a case is described in which modulation sections  102  and  112  execute BPSK modulation as shown in  FIG. 22 , but other modulation processing such as QPSK or 16 QAM may also be executed. 
     Transmission unit  2000  is provided in a base station, for example, which has a reception unit  200  as shown in  FIG. 2 . Transmission unit  2000  transmits signals with the frame configurations shown in  FIG. 3 . 
     Next, the configuration of a receiving apparatus will be described.  FIG. 21  shows the configuration of a reception unit of this embodiment that receives signals transmitted from transmission unit  2000 . Reception unit  2100  is provided in a communication terminal, for example. The difference between reception unit  2100  of this embodiment and reception unit  400  in  FIG. 4  is that reception unit  2100  has an eigenvalue based coefficient calculation section  2101 , soft decision value calculation sections  2102  and  2104 , error correction decoding sections  2103  and  2105 , and reception level based coefficient calculation section  2106 . The rest of the configuration is the same as that of reception unit  400  in  FIG. 4 , and therefore a description thereof is omitted here. 
     Eigenvalue based coefficient calculation section  2101  has channel fluctuation estimation information  427  as input, and outputs a coefficient  2110  found from an eigenvalue. Specifically, as also described in Embodiment 1, channel fluctuation h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) estimates are input as channel fluctuation estimation information  427 , Equation (3) channel fluctuation matrix eigenvalue calculation is performed with these estimates as elements, and coefficient  2110  is found based on the value with the smallest power among the eigenvalue powers. That is to say, coefficient  2110  is found by performing the same calculation as performed by eigenvalue based coefficient calculation section  214  ( FIG. 2 ) described in Embodiment 1, and this coefficient  2110  is sent to soft decision value calculation sections  2102  and  2104 . 
     Reception level based coefficient calculation section  2106  has received quadrature baseband signals  406  and  416  as input, calculates coefficients  2115  and  2116  based on received quadrature baseband signals  406  and  416 , and sends these coefficients  2115  and  2116  to soft decision value calculation sections  2102  and  2104  respectively. Specifically, spread signal A reception level based coefficient  2115  is found based on the reception level of the despread signal (received quadrature baseband signal) for spread signal A obtained by despreading section  405  and  415  respectively, and this coefficient  2115  is sent to soft decision value calculation section  2102 . Similarly, spread signal B reception level based coefficient  2116  is found based on the reception level of the despread signal (received quadrature baseband signal) for spread signal B obtained by despreading section  405  and  415  respectively, and this coefficient  2116  is sent to soft decision value calculation section  2104 . 
     Soft decision value calculation section  2102  has spread signal A received quadrature baseband signal  422 , coefficient  2115  found from the reception levels, and coefficient  2110  found from the eigenvalues as input, obtains a soft decision value by multiplying spread signal A received quadrature baseband signal  422  by coefficients  2115  and  2110 , and outputs this soft decision value as soft decision value signal  2111 . Error correction decoding section  2103  has soft decision value signal  2111  as input, and obtains and outputs received digital signal  2112  that has been error correction decoded by executing error correction decoding processing on soft decision value signal  2111 . 
     Soft decision value calculation section  2104  has spread signal B received quadrature baseband signal  423 , coefficient  2116  found from the reception levels, and coefficient  2110  found from the eigenvalues as input, obtains a soft decision value by multiplying spread signal B received quadrature baseband signal  423  by coefficients  2116  and  2110 , and outputs this soft decision value as soft decision value signal  2113 . Error correction decoding section  2105  has soft decision value signal  2113  as input, and obtains and outputs received digital signal  2114  that has been error correction decoded by executing error correction decoding processing on soft decision value signal  2113 . 
     It is assumed that a receiving apparatus (communication terminal) according to this embodiment has a transmission unit  500  as shown in  FIG. 5  in addition to reception unit  2100  shown in  FIG. 21 , and transmits signals with the frame configuration shown in  FIG. 6  from transmission unit  500 . 
     The operation of a transmitting apparatus and receiving apparatus according to this embodiment will now be described in detail. In this embodiment the receiving apparatus has special features, and therefore the operation of the receiving apparatus will be described in particular detail. The description will focus on operations differing from those in Embodiment 1, omitting operations that are the same as those in Embodiment 1. 
     Reception unit  2100  executes radio signal processing, despreading processing, channel fluctuation estimation processing for each spread signal, and so forth, on signals received at antennas  401  and  402 , then performs Equation (3) inverse matrix computation in signal processing section  421 , and obtains spread signal A received quadrature baseband signal  422  and spread signal B received quadrature baseband signal  423 . 
     It is here assumed that reception unit  2100  receives a BPSK modulated signal with a signal point arrangement as shown in  FIG. 22 . When coordinates of two points in the IQ plane are normalized by (+1.0,0.0) and (−1.0,0.0) in BPSK modulation, the soft decision value of received quadrature baseband signal R′(t) in the example shown in  FIG. 22  is +0.6 as shown in  FIG. 23 . 
     Important points concerning operation in this embodiment are that weighting is performed in soft decision value calculation sections  2102  and  2104  on a soft decision value obtained from a received quadrature baseband signal as described above, and more particularly that weighting is performed using coefficients found from eigenvalues. 
     To be specific, firstly, matrix eigenvalues shown in Equation (3) are found by eigenvalue based coefficient calculation section  2101  using channel fluctuation estimation information  427 —that is, estimated h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t )—generated by channel fluctuation information generation section  426 , and coefficient D(t)  2110  is calculated from the value with the smallest power among the eigenvalues. 
     On the other hand, in reception level based coefficient calculation section  2106 , reception level based coefficients Ca(t)  2115  and Cb(t)  2116  are obtained from the R 1 ( t ) and R 2 ( t ) reception levels (in this embodiment, R 1 ( t ) and R 2 ( t ) are despread signals). 
     Using coefficients D(t) and Ca(t) obtained as described above and received quadrature baseband signal R′a(t)  422 , received signal soft decision value Sa(t)  2111  is calculated by soft decision value calculation section  2102  using the following equation. 
       [Equation 5] 
         S   a ( t )= C   a ( t )× D ( t )× R′   a ( t )  (5)
 
     Similarly, using coefficients D(t) and Cb(t) obtained as described above and received quadrature baseband signal R′b(t)  423 , received signal soft decision value Sb(t)  2113  is calculated by soft decision value calculation section  2104  using the following equation. 
       [Equation 6] 
         S   b ( t )= C   b ( t )× D ( t )× R′   b ( t )  (6)
 
     In error correction decoding section  2103 , error correction decoding processing is performed using soft decision value Sa(t)  2111  obtained as described above. Similarly, in error correction decoding section  2105 , error correction decoding processing is performed using soft decision value Sb(t)  2113  obtained as described above. 
     Here, coefficients Ca(t)×D(t) and Cb(t)×D(t) for weighting used by soft decision value calculation sections  2102  and  2104  indicate the effective received field strength obtained by multiplying the received field strength actually received by an efficiency coefficient. Performing multiplication by this coefficient enables reception characteristics to be improved. 
     In this embodiment, convolutional coding is executed as error correction coding, and therefore maximum likelihood decoding such as Viterbi decoding is used. As regards the way in which a soft decision value is used in maximum likelihood decoding, methods previously disclosed in various documents include, for example, a method whereby the Euclidian distance between a soft decision value and each signal point is calculated and used, and a method whereby a metric value is calculated based on probability density distribution characteristics. In this embodiment, it is assumed, as an example, that the square Euclidian distance is calculated. That is to say, in the example shown in  FIG. 23 , likelihood metric values M 0  and M 1  from each signal point are calculated as shown in following Equation (7) and Equation (8) respectively. By this means, received digital signals  2112  and  2113  decoded by Viterbi coding are obtained. 
       [Equation 7] 
         M   0 ( t )=(+0.6−(−1.0)) 2 =2.56  (7)
 
       [Equation 8] 
         M   1 ( t )=(+0.6−(+1.0)) 2 =0.16  (8)
 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas a plurality of modulated signals transmitted from a plurality of antennas, by weighting a soft decision value using a coefficient D(t) based on the minimum value of eigenvalues calculated from channel fluctuation estimation results when performing error correction decoding using a received baseband signal obtained by separation, it is possible to give a soft decision value an appropriate likelihood based on effective reception power, enabling receive data error rate characteristics to be improved. 
     In this embodiment, in calculating reception level based coefficients, reception level based coefficient calculation section  2106  ( FIG. 21 ) finds spread signal A reception level based coefficient  2115  and spread signal B reception level based coefficient  2116  based on the output from despreading sections  405  and  415 , but coefficients  2115  and  2116  may also be found using channel estimation information h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) obtained by channel fluctuation estimation sections  407 ,  409 ,  417 , and  419 , in the same way as with reception power based coefficient calculation section  211  in  FIG. 2  described in Embodiment 1, or may be found from the RSSI (Received Signal Strength Indicator) of the received signal received from each antenna. This also applies to other embodiments in which processing is performed that uses reception level based coefficients. 
     Also, in this embodiment a case has been described in which soft decision value weighting is performed using reception level based coefficients  2115  and  2116  in addition to eigenvalue based coefficient  2110 , but soft decision value weighting may also be performed using only an eigenvalue based coefficient. 
     Moreover, the configuration of the transmission unit of a base station is not limited to that shown in  FIG. 20 . For example, transmission power modification sections  108  and  118  are not essential, and a configuration may be used whereby modulated signals  107  and  117  are supplied directly to antennas  110  and  120 . 
     Furthermore, a function that performs error detection coding, an interleaving function that switches around the signal order, a puncturing function that reduces redundancy by eliminating some signals, or the like, may be provided before or after error correction coding sections  2001  and  2002 , as necessary, without affecting the present invention. This also applies to other embodiments that have error correction coding sections. 
     Also, in this embodiment a case has been described in which error correction coding sections  2001  and  2002  perform error correction coding processing using convolutional code, but the error correction code used in error correction coding processing is not limited to convolutional code, and other code may be used as long as it is error correction code that allows decoding processing using a soft decision value during decoding. In this case, error correction decoding sections  2103  and  2105  of reception unit  2100  should perform decoding processing corresponding to the relevant coding. Moreover, a configuration may be used in which error correction coding sections  2001  and  2002  are combined into a single error correction coding section, and a coded signal is separated into two signals that are supplied to modulation section  102  and modulation section  112  respectively. In this case, error correction decoding sections  2103  and  2105  of reception unit  2100  can also be combined into a single decoding processing section. These comments also apply to other embodiments that have error correction coding processing sections. 
     Furthermore, in this embodiment a reception unit  2100  with the configuration shown in  FIG. 21  has been described as an example, but it is essential only that soft decision value weighting be performed using a coefficient based on the smallest value of eigenvalues calculated from channel fluctuation estimation results, and the reception unit configuration is not limited to that shown in  FIG. 21 , but may also be as shown in  FIG. 24 , for example. 
     The difference between reception unit  2400  in  FIG. 24  and reception unit  2100  in  FIG. 21  is that, whereas reception unit  2100  in  FIG. 21  performs signal separation processing by means of an inverse matrix computation by signal processing section  421 , reception unit  2400  in  FIG. 24  performs MLD (Maximum Likelihood Detection) by means of soft decision value calculation section  2401 , and then in error correction decoding section  2403  separates soft decision value signal  2402  into spread signal A received digital signal  2404  and spread signal B received digital signal  2405 . In performing this MLD, use of eigenvalue based coefficient  2110  enables receive data error rate characteristics to be improved in the same way as in the above-described embodiment. 
     By way of example, a case will here be described in which signals that have undergone QPSK modulation by modulation sections  102  and  112  of transmission unit  2000  shown in  FIG. 20  are demodulated by performing MLD in reception unit  2400  in  FIG. 24 . 
     Soft decision value calculation section  2401 , having received quadrature baseband signals  406  and  416 , channel fluctuation estimation information  408 ,  410 ,  418 , and  420 , reception level based coefficients  2115  and  2116 , and eigenvalue based coefficient  2110  as input, first calculates received quadrature baseband signal  406  and  416  candidate signal point positions (the present example assumes QPSK, with four candidate signal points provided per channel, so there are total 4×4=16 candidate signal point positions) using channel fluctuation estimation information  408 ,  410 ,  418 , and  420 , thereafter finds the signal point distance between these candidate points and reception point, and outputs that signal point distance weighted by reception level based coefficients  2115  and  2116  and eigenvalue based coefficient  2110  as soft decision value signal  2402 . 
     The process will now be described in detail.  FIG. 25(   a ) shows signal point position  2501  of received quadrature baseband signal  406  (the signal received by antenna  401  (antenna  1 )) and candidate signal point positions, and  FIG. 25(   b ) shows signal point position  2502  of received quadrature baseband signal  416  (the signal received by antenna  411  (antenna  2 )) and candidate signal point positions. 
     Soft decision value calculation section  2401  establishes candidate signal points of 4 transmit bits ( 0000 ), ( 0001 ), . . . , ( 1111 ) from spread signal A channel fluctuation estimation signal  408  and spread signal B channel fluctuation estimation signal  410  as shown in  FIG. 25(   a ). Then the distance between signal point  2501  of received quadrature baseband signal  406  and each candidate signal point is found. In fact, the square (power value) of the signal point distance is found. Here, the squares of the signal point distances between 4 transmit bits ( 0000 ), ( 0001 ), . . . , ( 1111 ) and reception point  2501  are denoted by x 0000 ( t ), x 0001 ( t ), x 0010 ( t ), and x 1111 ( t ) respectively. 
     Similarly, soft decision value calculation section  2401  establishes candidate signal points of 4 transmit bits ( 0000 ), ( 0001 ), . . . , ( 1111 ) from spread signal A channel fluctuation estimation signal  418  and spread signal B channel fluctuation estimation signal  420  as shown in  FIG. 25(   b ). Then the distance between signal point  2502  of received quadrature baseband signal  416  and each candidate signal point is found. In fact, the square (power value) of the signal point distance is found. Here, the squares of the signal point distances between 4 transmit bits ( 0000 ), ( 0001 ), . . . , ( 1111 ) and reception point  2502  are denoted by y 0000 ( t ), y 0001 ( t ), y 0010 ( t ), and y 1111 ( t ) respectively. 
     Soft decision value calculation section  2401  then performs soft decision value weighting using eigenvalue based coefficient  2110  and reception level based coefficients  2115  and  2116 . To be specific, calculation is performed as follows: weighted soft decision value z 0000 ( t )=Ca(t)D(t) {x 0000 ( t )+y 0000 ( t )}. z 0001 ( t ), z 0010 ( t ), . . . , z 1111 ( t ) are found in the same way. Ca(t) may be replaced by Cb(t). Soft decision value calculation section  2401  outputs z 0001 ( t ), z 0010 ( t ), z 1111 ( t ) weighted in this way as soft decision value signal  2402 . 
     By performing error correction decoding of soft decision value signal  2402  that has undergone MLD processing and eigenvalue based weighting processing in this way, error correction coding section  2403  obtains spread signal A received digital signal  2404  and spread signal B received digital signal  2405 , and outputs these signals. 
     Embodiment 8 
     In this embodiment, a case is described in which the processing described in Embodiment 7 is applied to OFDM communications. A special feature of this embodiment is that the processing whereby soft decision values are weighted using eigenvalue based coefficients calculated from channel fluctuation estimation results is performed for each subcarrier. 
       FIG. 26  shows a sample configuration of the transmission unit of a transmitting apparatus according to this embodiment. Parts in  FIG. 26  corresponding to parts in  FIG. 10  are assigned the same codes as in  FIG. 10 , and descriptions of parts previously described using  FIG. 10  are omitted. 
     The difference between transmission unit  2600  in  FIG. 26  and transmission unit  1000  in  FIG. 10  is that transmission unit  2600  has error correction coding sections  2601  and  2603 , which execute error correction coding processing on transmit digital signals  101  and  111  using convolutional code, and send error correction coded signals  2602  and  2604  to modulation sections  102  and  112 . By this means, transmission unit  2600  performs OFDM processing of error correction coded data, enabling transmit data to be coded in the frequency axis direction. 
     Transmission unit  2600  is provided in a base station, for example, which has a reception unit  200  as shown in  FIG. 2 . Transmission unit  2600  transmits signals with the frame configurations shown in  FIG. 11 . 
       FIG. 27  shows the configuration of a reception unit of this embodiment that receives signals transmitted from transmission unit  2600 . Reception unit  2700  is broadly configured as a combination of reception unit  1200  shown in  FIG. 12  and reception unit  2100  shown in  FIG. 21 , and therefore descriptions of previously described parts in  FIG. 12  and  FIG. 21  are omitted here, and only parts specific to this embodiment are described. Parts in  FIG. 27  corresponding to parts in  FIG. 12  are assigned the same codes as in  FIG. 12 . 
     Channel fluctuation estimation sections  1207 ,  1209 ,  1217 , and  1219  estimate channel fluctuation on a subcarrier-by-subcarrier basis based on estimation symbols arranged in each subcarrier. Channel fluctuation information generation section  2703  and eigenvalue based coefficient calculation section  2705  find eigenvalue based coefficient  2706  for each subcarrier by performing the same processing as in channel fluctuation information generation section  426  and eigenvalue based coefficient calculation section  2101  in  FIG. 21  on a subcarrier-by-subcarrier basis, and send eigenvalue based coefficient  2706  to soft decision value calculation sections  2707  and  2711 . 
     Reception level based coefficient calculation section  2701  has output signals  1204  and  1214  from radio sections  1203  and  1213 , output signals  1206  and  1216  from discrete Fourier transform sections (dft&#39;s)  1205  and  1215 , and output signals  1208 ,  1210 ,  1218 , and  1220  from channel fluctuation estimation sections  1207 ,  1209 ,  1217 , and  1219  as input, and using some or all of these, finds reception level based coefficient  2702  for each subcarrier, and sends this coefficient  2702  to soft decision value calculation sections  2707  and  2711 . 
     Soft decision value calculation sections  2707  and  2711  weight input channel A received quadrature baseband signal group  1222  and channel B received quadrature baseband signal group  1223  by means of eigenvalue based coefficient  2706  and reception level based coefficient  2702 , and output soft decision value signals  2708  and  2712 . Here, soft decision value calculation sections  2707  and  2711  perform the same kind of weighting processing as described for soft decision value calculation sections  2102  and  2104  in  FIG. 21  for each subcarrier. That is to say, different weighting processing is performed for each subcarrier using the same subcarrier received quadrature baseband signal, eigenvalue based coefficient, and reception level based coefficient. 
     In this way, soft decision value signals  2708  and  2712  weighted on a subcarrier-by-subcarrier basis are obtained, these soft decision value signals  2708  and  2712  undergo error correction decoding processing by error correction decoding sections  2709  and  2713 , and received digital signals  2710  and  2714  are obtained. 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas a plurality of OFDM modulated signals transmitted from a plurality of antennas, by performing processing whereby soft decision values are weighted using a coefficient based on an eigenvalue calculated from channel fluctuation estimation results on a subcarrier-by-subcarrier basis, it is possible to give a soft decision value an appropriate likelihood based on per-subcarrier effective reception power, enabling receive data error rate characteristics to be improved, even when per-subcarrier effective reception power varies due to frequency selective fading, etc. 
     In this embodiment a reception unit  2700  with the configuration shown in  FIG. 27  has been described as an example, but it is essential only that per-subcarrier soft decision value weighting be performed using a coefficient based on the smallest value of per-subcarrier eigenvalues calculated from per-subcarrier channel fluctuation estimation results, and the reception unit configuration is not limited to that shown in  FIG. 27 , but may also be as shown in  FIG. 28 , for example. 
     The difference between reception unit  2800  in  FIG. 28  and reception unit  2700  in  FIG. 27  is that, whereas reception unit  2700  in  FIG. 27  performs signal separation processing by means of an inverse matrix computation by signal processing section  1221 , reception unit  2800  in  FIG. 28  performs MLD (Maximum Likelihood Detection) by means of soft decision value calculation section  2801 , and then in error correction decoding section  2803  separates soft decision value signal  2802  into received digital signal  2804  and received digital signal  2805 . 
     As MLD processing has been described in Embodiment 7 using  FIG. 24 , a description thereof is omitted here. However, the difference between above-described soft decision value calculation section  2401  in  FIG. 24  and soft decision value calculation section  2801  of this embodiment in  FIG. 28  is that soft decision value calculation section  2801  performs the same kind of processing as soft decision value calculation section  2401  on a subcarrier-by-subcarrier basis. That is to say, soft decision value calculation section  2801  performs processing on a subcarrier-by-subcarrier basis to calculate all of candidate signal point positions on received quadrature baseband signal group  1206  and  1216  using channel fluctuation estimation information  1208 ,  1210 ,  1218 , and  1220 , then finds the signal point distance between the candidate points and reception point on a subcarrier-by-subcarrier basis, and outputs that signal point distance weighted by reception level based coefficient  2702  and eigenvalue based coefficient  2760  as soft decision value signal  2802  on a subcarrier-by-subcarrier basis. In other words, per-subcarrier soft decision values are output as soft decision value signal  2802 . 
     Embodiment 9 
     A special feature of this embodiment is that, in contrast to Embodiment 7, error correction coding processing is not performed individually on data transmitted from each antenna, but instead, data is supplied to each antenna after undergoing error correction coding processing by a single error correction coding section. As a result, when MLD (Maximum Likelihood Detection) processing and error correction decoding processing are performed on the receiving side, single-system error correction code is input to the MLD processing section and error correction decoding section, enabling data with improved error rate characteristics to be obtained. 
       FIG. 29 , in which parts corresponding to parts in  FIG. 20  are assigned the same codes as in  FIG. 20 , shows the configuration of a transmission unit  2900  of this embodiment. The difference between transmission unit  2900  of this embodiment and transmission unit  2000  in  FIG. 20  is that, whereas transmission unit  2000  has error correction coding sections  2001  and  2002  for antennas  110  and  120  respectively and performs error correction coding processing of transmit digital signals  101  and  111  individually for antennas  110  and  120 , in transmission unit  2900  error correction coding section  2902  first performs error correction processing on transmit digital signal  2901  and then splits the data into error correction coded data  2903  and  2904 , and supplies error correction coded data  2903  and  2904  to modulation sections  102  and  112  respectively. 
       FIG. 30 , in which parts corresponding to parts in  FIG. 24  are assigned the same codes as in  FIG. 24 , shows the configuration of a reception unit  3000  of this embodiment. Reception unit  3000  receives signals transmitted from transmission unit  2900 . That is to say, reception unit  3000  receives signals that have undergone error correction coding by the single error correction coding section  2902 . As a result, soft decision value calculation section  2401  and error correction decoding section  3001  perform error correct ion decoding processing by calculating a single-system error correction coded signal soft decision value, and thus error correction capability is improved compared with a case where error correction decoding processing is performed by calculating soft decision values separately for multi-system error correction coded signals (for example, compared with reception unit  2400  in  FIG. 24 ). By this means, a received digital signal  3002  with improved error rate characteristics can be obtained. 
     Thus, according to this embodiment, when transmit data undergoes error correction coding processing and is transmitted from a plurality of antennas, transmit data is error correction coded by a single error correction coding section  2902 , in contrast to the configuration in Embodiment 7, making it possible to improve error correction capability when MLD processing and error correction decoding processing are performed on the receiving side, and enabling receive data with greatly improved error rate characteristics to be obtained. 
     Embodiment 10 
     In this embodiment, a case is described in which the special feature of Embodiment 9 is applied to OFDM communications. 
       FIG. 31  shows a sample configuration of the transmission unit of a transmitting apparatus according to this embodiment. Parts in  FIG. 31  corresponding to parts in  FIG. 26  are assigned the same codes as in  FIG. 26 . The difference between transmission unit  3100  of this embodiment and transmission unit  2600  in  FIG. 26  is that, whereas transmission unit  2600  has error correction coding sections  2601  and  2602  for antennas  110  and  120  respectively and performs error correction coding processing of transmit digital signals  101  and  111  individually for antennas  110  and  120 , in transmission unit  3100  error correction coding section  3102  first performs error correction processing on transmit digital signal  3101  and then splits the data into error correction coded data  3103  and  3104 , and supplies error correction coded data  3103  and  3104  to modulation sections  102  and  112  respectively. 
       FIG. 32 , in which parts corresponding to parts in  FIG. 28  are assigned the same codes as in  FIG. 28 , shows the configuration of a reception unit  3200  of this embodiment. Reception unit  3200  receives signals transmitted from transmission unit  3100 . That is to say, reception unit  3200  receives signals that have undergone error correction coding by the single error correction coding section  3102 . As a result, soft decision value calculation section  2801  and error correction decoding section  3201  perform error correct ion decoding processing by calculating a single-system error correction coded signal soft decision value, and thus error correction capability is improved compared with a case where error correction decoding processing is performed by calculating soft decision values separately for multi-system error correction coded signals (for example, compared with reception unit  2800  in  FIG. 28 ). By this means, a received digital signal  3202  with improved error rate characteristics can be obtained. 
     Thus, according to this embodiment, when transmit data undergoes error correction coding processing and is transmitted from a plurality of antennas, transmit data is error correction coded by a single error correction coding section  3102 , in contrast to the configuration in Embodiment 8, making it possible to improve error correction capability when MLD processing and error correction decoding processing are performed on the receiving side, and enabling receive data with greatly improved error rate characteristics to be obtained. 
     Embodiment 11 
     A special feature of this embodiment is that, in a receiving apparatus that performs demodulation processing using channel fluctuation matrix eigenvalues, a reception level control section is provided that detects the signal level of each antenna received signal and makes the signal levels of the antenna received signals equal. 
       FIG. 33 , in which parts corresponding to parts in  FIG. 21  are assigned the same codes as in  FIG. 21 , shows the configuration of a reception unit  3300  of this embodiment. Except for the provision of a reception level control section  3301 , reception unit  3300  has the same configuration as reception unit  2100  in  FIG. 21 . 
     Reception level control section  3301  has received quadrature baseband signals  404  and  414  as input, detects the signal levels of these received quadrature baseband signals  404  and  414 , and sends gain control signals  3302  and  3303  for equalizing the signal levels of received quadrature baseband signals  404  and  414  to radio sections  403  and  413 . Radio sections  403  and  413  change the amplifier gain based on gain control signals  3302  and  3303 . 
     The operation of reception unit  3300  of this embodiment will now be described. Reception unit  3300  performs control by means of reception level control section  3301  so that the levels of the received signals received by antennas  401  and  411  become equal—that is to say, so that the output levels of received quadrature baseband signals  404  and  414  output from radio sections  403  and  413  respectively become equal. 
     For example, if a −40 dBm signal is received by antennas  401  and  411 , control is performed so that the voltages of received quadrature baseband signals  404  and  414  are 2 V. On the other hand, if a −40 dBm signal is received by antenna  401  and a −46 dBm signal is received by antenna  411 , control is not performed so that the voltages of received quadrature baseband signals  404  and  414  are both 2 V, but instead, control is performed so that the voltage of received quadrature baseband signal  404  is 2 V and the voltage of received quadrature baseband signal  414  is 1 V. In this way, the signal levels of received quadrature baseband signals  404  and  414  are made equal. 
     Making the signal levels from the antennas equal in this way greatly improves demodulation precision when performing demodulation using channel fluctuation matrix eigenvalues, because the closer the received signal levels of the antennas, the greater is the significance of a channel fluctuation matrix eigenvalue. When the signal level of each antenna received signal is controlled separately and control is performed so that received quadrature baseband signals  404  and  414  have the same voltage, the significance of an eigenvalue as an effective reception power index decreases. 
     In those of the above-described embodiments in which reception level based coefficients are used together with eigenvalue based coefficients, and demodulation is performed with these coefficients as effective reception power indices, even if the signal levels of antenna received signals are different, the same effect can be obtained as by equalizing reception levels, as in this embodiment, if eigenvalues are corrected by reflecting this difference of signal levels in a reception level coefficient. 
     Thus, according to this embodiment, in a receiving apparatus that performs demodulation processing using channel fluctuation matrix eigenvalues, by detecting the signal level of each antenna received signal and equalizing the signal levels of the antenna received signals, the value of an eigenvalue can be made a much more appropriate value for use as an effective reception power index, and receive data with greatly improved error rate characteristics can be obtained. 
     Control of the signal level of each antenna received signal is not limited to application to reception unit  3300  with the configuration shown in  FIG. 33 , but can be widely applied to cases where demodulation processing is performed using eigenvalues. 
     Embodiment 12 
     In this embodiment, a case is described in which the special feature of Embodiment 11 is applied to OFDM communications. 
       FIG. 34 , in which parts corresponding to parts in  FIG. 27  are assigned the same codes as in  FIG. 27 , shows the configuration of a reception unit  3400  of this embodiment. Except for the provision of a reception level control section  3401 , reception unit  3400  has a similar configuration to reception unit  2700  in  FIG. 27 . 
     Reception level control section  3401  has received quadrature baseband signals  1204  and  1214  as input, detects the signal levels of these received quadrature baseband signals  1204  and  1214 , and sends gain control signals  3402  and  3403  for equalizing the signal levels of received quadrature baseband signals  1204  and  1214  to radio sections  1203  and  1213 . Radio sections  1203  and  1213  change the amplifier gain based on gain control signals  3402  and  3403 . 
     By thus performing control in such a way that makes the signal levels of antenna received signals equal, the signal levels between the subcarriers corresponding to post-Fourier-transform signals  1206  and  1216  can also be made virtually equal. By this means, when modulation is performed using channel fluctuation matrix eigenvalues on a subcarrier-by-subcarrier basis, eigenvalues for each subcarrier can be made to reflect accurately per-subcarrier effective reception power. 
     Thus, according to this embodiment, in a receiving apparatus that performs demodulation processing using channel fluctuation matrix eigenvalues on a subcarrier-by-subcarrier basis, by detecting the signal level of each antenna received signal and equalizing the signal levels of the antenna received signals, the value of a per-subcarrier eigenvalue can be made a much more appropriate value for use as an effective reception power index, and OFDM receive data with greatly improved error rate characteristics can be obtained. 
     Embodiment 13 
     In this embodiment, it is proposed that space-time coded modulated signals be transmitted from a plurality of antennas, and received signals be demodulated on the receiving side using channel fluctuation matrix eigenvalues. In this embodiment, in particular, a receiving antenna is selected using channel fluctuation matrix eigenvalues, and received signal demodulation is performed using only the space-time coded signal received by the selected receiving antenna. 
     Space-time coding is a known technology, and is described, for example, in “Space-Time Block Codes from Orthogonal Design” IEEE Transactions on Information Theory, pp. 1456-1467, vol. 45, no. 5, July 1999. 
     An overview of space-time coding will be given using  FIG. 35  and  FIG. 36 . In a communication system that uses space-time coding, transmit signal A shown in  FIG. 35  is transmitted from a transmitting antenna  3601 , and at the same time, transmit signal B shown in  FIG. 35  is transmitted from a transmitting antenna  3602 . When this is done, transmit signal A and transmit signal B transmitted from transmitting antennas  3601  and  3602  are subjected to channel fluctuations h 1 ( t ) and h 2 ( t ) respectively, and are received by a receiving antenna  3603 . 
     In  FIG. 35 , reference numerals  3501  and  3504  indicate radio wave propagation environment symbols, and reference numerals  3502 ,  3503 ,  3505 , and  3506  indicate coded symbol groups. Also, S 1  and S 2  are assumed to be different signals, and signal S 1  is sent in symbol group  3502 , signal −S 2 *, which is the negative complex conjugate of signal S 2 , is sent in symbol group  3503 , signal S 2  is sent in symbol group  3505 , and signal S 1 *, which is the complex conjugate of signal S 1 , is sent in symbol group  3506 . An asterisk (*) here indicates a complex conjugate. 
     The relationship between signals S 1  and S 2  transmitted from transmitting antennas  3601  and  3602 , and signals R 1  and R 2  received by receiving antenna  3603  can then be expressed by the following equation. 
     
       
         
           
             
               
                 
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     In Equation (9), R 1  is the received signal when symbol group  3502  and symbol group  3505  in  FIG. 35  are received, and R 2  is the received signal when symbol group  3503  and symbol group  3506  in  FIG. 35  are received. 
     As can be seen from Equation (9), if this kind of space-time coding technology is used, transmit signals S 1  and S 2  to be found can be obtained by received signal maximal-ratio combining, and therefore a transmit signal can be estimated with good precision from a received signal . This concludes the overview of space-time coding technology. 
     The configuration of this embodiment will now be described.  FIG. 37 , in which parts corresponding to parts in  FIG. 29  described in Embodiment 9 are assigned the same codes as in  FIG. 29 , shows the configuration of a transmission unit  3700  of a transmitting apparatus according to this embodiment. The difference between transmission unit  2900  in  FIG. 29  and transmission unit  3700  of this embodiment is that error correction coding section  3701  of transmission unit  3700  performs space-time coding processing on transmit digital signal  2901  and outputs the resulting signals. That is to say, error correction coding section  3701  performs coding processing so that the relationship between error correction coded signal  2903  and error correction coded signal  2904  is of the same kind as between transmit signal A and transmit signal B in  FIG. 35 . By this means, space-time coded signals are transmitted from antennas  110  and  120  of transmission unit  3700 . 
       FIG. 38  shows a configuration of a reception unit  3800  that receives space-time coded signals transmitted from transmission unit  3700 . Parts in  FIG. 38  corresponding to those in  FIG. 14  described in Embodiment 3 are assigned the same codes as in  FIG. 14 . The differences between reception unit  1400  in  FIG. 14  and reception unit  3800  of this embodiment will be described here. 
     Antenna selection section  1411  of reception unit  1400  of Embodiment 3 creates two antenna received signal combinations from three antennas&#39; received signals  408 ,  410 ,  406 ,  418 ,  420 ,  416 ,  1408 ,  1410 , and  1406  containing channel estimates, finds an eigenvalue for each combination, and selects two antennas&#39; received signals of the combination for which the eigenvalue minimum power is greatest, and outputs these as selected signals  1412 ,  1413 ,  1414 ,  1415 ,  1416 , and  1417 . 
     In contrast to this, antenna selection section  3801  of reception unit  3800  of this embodiment finds an eigenvalue for each antenna received signal (that is, finds the Equation (9) eigenvalue for each antenna received signal) from three antennas&#39; received signals  408 ,  410 ,  406 ,  418 ,  420 ,  416 ,  1408 ,  1410 , and  1406  containing channel estimates, selects one antennas&#39; received signals for which the eigenvalue minimum power is greatest, and outputs these as selected signals  3802 ,  3803 , and  3804 . The reason why it is possible to find an eigenvalue for the received signals of each antenna in this way is that a signal received at each antenna is a space-time coded signal and a channel estimation matrix as shown in Equation (9) is obtained only for one antenna&#39;s received signals. 
     Also, signal processing section  421  of reception unit  1400  of Embodiment 3 obtains two received quadrature baseband signals  422  and  423  by separating input two antennas&#39; received signals  1412 ,  1413 ,  1414 ,  1415 ,  1416 , and  1417  by means of the inverse matrix computation of Equation (3). 
     In contrast to this, signal processing section  3805  of reception unit  3800  of this embodiment obtains S 1  and S 2  received digital signal  3806  by performing maximal-ratio combining of input one antennas&#39; received signals based on Equation (9). 
       FIG. 39  shows the configuration of antenna selection section  3801 . Antenna selection section  3801  has an eigenvalue calculation section  3901  and a signal selection section  3903 . Eigenvalue calculation section  3901  has channel fluctuations  408  and  410 ,  418  and  420 , and  1408  and  1410 , obtained from the received signals of each antenna, as input. Eigenvalue calculation section  3901  finds an Equation (9) eigenvalue using channel fluctuations  408  and  410 . Similarly, eigenvalue calculation section  3901  finds an Equation (9) eigenvalue using channel fluctuations  418  and  420 , and finds an Equation (9) eigenvalue using channel fluctuations  1408  and  1410 . The eigenvalue minimum powers are then compared, the antenna for which the eigenvalue minimum power is greatest is detected, and a control signal  3902  indicating that antenna is sent to signal selection section  3903 . 
     Signal selection section  3903  outputs signals corresponding to the antenna indicate by control signal  3902  from among signals  408 ,  410 , and  406  obtained from the antenna  401  received signal, signals  418 ,  420 , and  416  obtained from the antenna  411  received signal, and signals  1408 ,  1410 , and  1406  obtained from the antenna  1401  received signal, as selected signals  3602 ,  3603 , and  3604 . 
     The operation of reception unit  3800  of this embodiment will now be described. Reception unit  3800  receives space-time coded signals, transmitted from receiving antennas  110  and  120  ( FIG. 37 ), by means of receiving antennas  401 ,  411 , and  1401 . Reception unit  3800  estimates channel fluctuation values h 1 ( t ) and h 2 ( t ) for each receiving antenna. 
     Reception unit  3800  then calculates the channel fluctuation matrix eigenvalue shown in Equation (9) for each receiving antenna from the channel fluctuation values of each receiving antenna by means of antenna selection section  3801 . Antenna selection section  3801  selects the antenna received signals for which the eigenvalue minimum power is greatest. By this means, the antenna received signals for which the effective reception power is greatest are selected. Reception unit  3800  then obtains receive data by demodulating the selected antenna received signals. 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas space-time coded signals transmitted from a plurality of antennas, by calculating channel fluctuation matrix eigenvalues of the space-time coded signals received by each antenna, selecting the antenna received signal for which the eigenvalue minimum power is greatest, and performing demodulation processing thereupon, it is possible to select the antenna received signal with the greatest effective reception power, enabling receive data with good error rate characteristics to be obtained. 
     In this embodiment, a case has been described in which the number of transmitting antennas is two, and the kind of space-time code shown in  FIG. 35  is used, but the number of transmitting antennas is not limited to two, and the space-time code is not limited to that shown in  FIG. 35 . 
     Embodiment 14 
     In this embodiment, a case is described in which, as in Embodiment 13, when space-time coded modulated signals are transmitted from a plurality of antennas, channel fluctuation matrix eigenvalues are found for each antenna&#39;s received signals on the receiving side, and the antenna received signals for which the eigenvalue minimum power is greatest are selected and undergo demodulation. However, in this embodiment, a case is described in which the special feature of Embodiment 13 is applied to OFDM communications. 
       FIG. 40  shows frame configurations when space-time code is OFDM modulated and transmitted. As can be seen by comparing  FIG. 40  with  FIG. 35 , space-time code is arranged in carrier  1  of the same frequency band. Mutually corresponding codes are also similarly arranged in other carriers. Such transmit signals A and B can be formed by replacing spreading sections  104  and  114  in  FIG. 37  with inverse discrete Fourier transform sections (idft&#39;s). 
     In a reception unit that receives signals with the kind of frames shown in  FIG. 40 , despreading sections  405 ,  415 , and  1405  in  FIG. 38  can be replaced by discrete Fourier transform sections (dft&#39;s), spread signal A channel fluctuation estimation sections  407 ,  417 , and  1407  can be replaced by channel A channel fluctuation estimation sections, and spread signal B channel fluctuation estimation sections  409 ,  419 , and  1409  can be replaced by channel B channel fluctuation estimation sections. It is assumed that the channel A channel fluctuation estimation sections estimate per-subcarrier channel fluctuation, and the channel B channel fluctuation estimation sections similarly estimate per-subcarrier channel fluctuation. 
     Antenna selection section  3801  can then calculate channel fluctuation matrix eigenvalues of space-time coded signals received at each antenna on a subcarrier-by-subcarrier basis, and select antenna received signals for which the eigenvalue minimum power is greatest on a subcarrier-by-subcarrier basis. 
     In this way, the antenna for which effective reception power is greatest can be selected on a subcarrier-by-subcarrier basis, enabling the optimal antenna to be selected for each subcarrier. As a result, error rate characteristics can be improved for all subcarriers. 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas space-time coded OFDM modulated signals transmitted from a plurality of antennas, by calculating on a subcarrier-by-subcarrier basis channel fluctuation matrix eigenvalues of the space-time coded signals received by each antenna, selecting on a subcarrier-by-subcarrier basis the antenna received signal for which the eigenvalue minimum power is greatest, and performing demodulation processing thereupon, it is possible to select on a subcarrier-by-subcarrier basis the antenna received signal with the greatest effective reception power, enabling receive data with good error rate characteristics to be obtained across all subcarriers. 
     In this embodiment, the kind of frame configuration shown in  FIG. 40  has been taken by way of example as the frame configuration used when space-time code is OFDM modulated and transmitted, but in a case where signals with the kind of frame configuration shown in  FIG. 41  are received by a plurality of antennas, also, as long as an antenna is selected based on channel fluctuation matrix eigenvalues for each receiving antenna, antenna received signals for which the effective reception power is greatest can be selected in the same way as in the above embodiment, enabling the error rate characteristics of receive data to be improved. The coding shown in  FIG. 41  is generally referred to as frequency-time coding as opposed to space-time coding. 
     That is to say, the eigenvalue-based receiving antenna selection method according to this embodiment is not limited to space-time coding, and the same kind of effect as in the above-described embodiment can also be obtained if the present invention is applied to space-frequency coding, or space-frequency-time coding in which space-time coding and space-frequency coding are performed simultaneously. 
     Embodiment 15 
     In above-described Embodiment 13, it was proposed that, when space-time coded received signals are received by a plurality of antennas, a receiving antenna be selected based on channel fluctuation matrix eigenvalues of each antenna&#39;s received signals (that is, only one receiving antenna be selected), and receive data be obtained by demodulating the signals obtained by the selected receiving antenna. 
     In contrast to this, in this embodiment a method and apparatus are proposed whereby, when space-time coded signals are received by a plurality of antennas, each antenna&#39;s received signals are weighted and combined based on channel fluctuation matrix eigenvalues of each antenna&#39;s received signals, and receive data is obtained by demodulating the weighted and combined received signals. 
     The eigenvalue-based antenna received signal weighting and combining method of this embodiment is similar to the combining method of above-described Embodiment 5. However, the combining method of this embodiment and the combining method of Embodiment 5 differ in the following respect. 
     In the combining method of Embodiment 5, a plurality of antenna received signal combinations are first created, a channel fluctuation matrix is created for each combination, and channel fluctuation matrix eigenvalues are calculated for each combination. Then, modulated signals are separated using the antenna received signals of each combination and the channel fluctuation matrix corresponding to that combination, and modulated signals separated in each combination are weighted and combined using the channel fluctuation estimation matrix eigenvalues used at the time of separation. 
     In contrast to this, in the combining method of this embodiment, a channel fluctuation matrix as shown in Equation (9) is created for each antenna&#39;s received signals, and an eigenvalue of the channel fluctuation matrix of each antenna&#39;s received signals is calculated. Each antenna&#39;s received signals are then weighted and combined based on these eigenvalues. In this kind of embodiment, antenna received signal combinations are not found as in Embodiment 5, but a channel fluctuation matrix is created individually for each antenna&#39;s received signals, and an eigenvalue is found individually for each antenna&#39;s received signals. This is possible because the received signals are space-time coded signals. 
       FIG. 42 , in which parts corresponding to parts in  FIG. 18  described in Embodiment 5 are assigned the same codes as in  FIG. 18 , shows the configuration of a reception unit  4200  according to this embodiment. The difference between reception unit  1800  in  FIG. 18  and reception unit  4200  of this embodiment lies in the configuration of signal processing section  4201 . Reception unit  4200  receives space-time coded signals as shown in  FIG. 35  transmitted from transmission unit  3700  shown in  FIG. 37 . 
       FIG. 43  shows the configuration of signal processing section  4201 . Signal processing section  4201  has an eigenvalue calculation section  4301  and a combining section  4303 . Eigenvalue calculation section  4301  has channel fluctuations  408  and  410 ,  418  and  420 , and  1408  and  1410 , obtained from the received signals of each antenna, as input. Eigenvalue calculation section  4301  finds an Equation (9) eigenvalue using channel fluctuations  408  and  410 . Similarly, eigenvalue calculation section  4301  finds an Equation (9) eigenvalue using channel fluctuations  418  and  420 , and finds an Equation (9) eigenvalue using channel fluctuations  1408  and  1410 . Eigenvalue calculation section  4301  then finds for each antenna the value with the minimum eigenvalue power from among the eigenvalues found for each antenna, and outputs the results as eigenvalue powers P 1 , P 2 , and P 3  of each antenna&#39;s received signals. That is to say, eigenvalue calculation section  4301  outputs eigenvalue powers P 1 , P 2 , and P 3  for each of antennas  1401 ,  411 , and  1401  as an eigenvalue estimation signal  4302 . 
     Combining section  4303  applies input signals  408 ,  410 , and  406  to Equation (9), and by performing Equation (9) inverse matrix computation, finds spread signal A received quadrature baseband signal Ra 1  and spread signal B received quadrature baseband signal Rb 1 . Similarly, combining section  4303  applies input signals  418 ,  420 , and  416  to Equation (9), and by performing Equation (9) inverse matrix computation, finds spread signal A received quadrature baseband signal Ra 2  and spread signal B received quadrature baseband signal Rb 2 . Similarly, combining section  4303  applies input signals  1408 ,  1410 , and  1406  to Equation (9), and by performing Equation (9) inverse matrix computation, finds spread signal A received quadrature baseband signal Ra 3  and spread signal B received quadrature baseband signal Rb 3 . 
     Next, combining section  4303  weights and combines these spread signal A received quadrature baseband signals Ra 1 , Ra 2 , and Ra 3 , and spread signal B received quadrature baseband signals Rb 1 , Rb 2 , and Rb 3 , using eigenvalue powers P 1 , P 2 , and P 3  of each antenna. Specifically, if spread signal A received quadrature baseband signal after weighting and combining  4202  is designated Ra and spread signal B received quadrature baseband signal after weighting and combining  4203  is designated Rb, then Ra and Rb are given by the following equations. 
     
       
         
           
             
               
                 
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     By weighting and combining each antenna&#39;s received signals according to eigenvalue power on an antenna-by-antenna basis in this way, accurate spread signal A and B received quadrature baseband signals can be obtained. This is because the channel fluctuation matrix eigenvalue power of each antenna&#39;s received signals is a value corresponding to the effective reception power of each antenna&#39;s received signals. 
     Spread signal A received quadrature baseband signal  4202  and spread signal B received quadrature baseband signal  4203  obtained by signal processing section  4201  are demodulated and decoded by demodulation units (not shown), to become received digital signals. 
     By this means, data can be demodulated using spread signal A and B received quadrature baseband signals  4202  and  4203  with large effective reception power, enabling received digital signals with improved error rate characteristics to be obtained. 
     Thus, according to this embodiment, in a receiving apparatus that receives at a plurality of antennas space-time coded signals transmitted from a plurality of antennas, by calculating channel fluctuation matrix eigenvalues of the space-time coded signals received by each antenna, and weighting and combining each antenna&#39;s received signals using per-antenna eigenvalue power, it is possible to obtain received signals with large effective reception power, enabling receive data with good error rate characteristics to be obtained. 
     In this embodiment a method has been described whereby channel fluctuation matrix eigenvalue power is used as a weighting coefficient, and received quadrature baseband signals are weighted and combined using this coefficient, but the present invention is not limited to this. 
     For example, channel fluctuation matrix eigenvalue power described in this embodiment may also be used as a weighting coefficient for MLD (Maximum Likelihood Detection) shown in “A simple transmit diversity technique for wireless communications” IEEE Journal on Select Areas in Communications, vol. 16, no. 8, October 1998. Use of channel fluctuation matrix eigenvalue power as a weighting coefficient in demodulation and decoding when performing MLD improves reception quality. This also applies to Embodiment 16 described below. 
     Embodiment 16 
     In this embodiment, a case is described in which, as in Embodiment 15, when space-time coded modulated signals are transmitted, channel fluctuation matrix eigenvalues are found for each antenna&#39;s received signals on the receiving side, and the antenna received signals for which the eigenvalue minimum power is greatest are selected and undergo demodulation. However, in this embodiment, a case is described in which the special feature of Embodiment 15 is applied to OFDM communications. 
     That is to say, a receiving apparatus of this embodiment receives signals with the frame configurations shown in  FIG. 40 . In the reception unit of a receiving apparatus of this embodiment, despreading sections  405 ,  415 , and  1405  in  FIG. 42  can be replaced by discrete Fourier transform sections (dft&#39;s), spread signal A channel fluctuation estimation sections  407 ,  417 , and  1407  can be replaced by channel A channel fluctuation estimation sections, and spread signal B channel fluctuation estimation sections  409 ,  419 , and  1409  can be replaced by channel B channel fluctuation estimation sections. It is assumed that the channel A channel fluctuation estimation sections estimate per-subcarrier channel fluctuation, and the channel B channel fluctuation estimation sections similarly estimate per-subcarrier channel fluctuation. 
     Signal processing section  4201  then calculates channel fluctuation matrix eigenvalues of space-time coded signals received at each antenna on a subcarrier-by-subcarrier basis, and performs weighting and combining using eigenvalue power described in Embodiment 15 as a weight coefficient on a subcarrier-by-subcarrier basis. 
     In this way, by performing combining of each antenna&#39;s received signals with eigenvalue power as a weight coefficient on a carrier-by-carrier basis, error rate characteristics can be improved across all carriers even when effective reception power differs greatly from carrier to carrier due to the effects of frequency selective fading, etc. 
     Thus, according to this embodiment, when space-time coded OFDM signals are received at a plurality of antennas, by performing processing whereby received signals obtained at each receiving antenna are weighted and combined based on channel fluctuation matrix eigenvalues as described in Embodiment 15, for each carrier, it is possible to implement a receiving apparatus that enables the error rate characteristics of received space-time coded OFDM signals to be improved across all carriers. 
     Embodiment 17 
     In this embodiment, receiving-side demodulation processing is described for a case where convolutional coded signals further undergo space-time block coding and are transmitted from a plurality of antennas. 
       FIG. 44 , in which parts corresponding to parts in  FIG. 1  described in Embodiment 1 are assigned the same codes as in  FIG. 1 , shows the configuration of a transmission unit  4400  of a transmitting apparatus of this embodiment. Error correction coding sections  4401  and  4405  of transmission unit  4400  have digital signals  101  and  111  as input respectively, execute convolutional coding, for example, and send coded digital signals  4402  and  4406  to a space-time block coding section  4403 . 
     Space-time block coding section  4403  has coded digital signals  4402  and  4406  as input, and by executing space-time block coding as shown in Equation (9) on these coded digital signals  4402  and  4406 , outputs modulated signal A transmit digital signal  4404  (corresponding to transmit signal A in  FIG. 35 ) and modulated signal B transmit digital signal  4407  (corresponding to transmit signal B in  FIG. 35 ) with the frame configurations shown in  FIG. 35 . 
     The kind of space-time block coding method in Equation (9) is shown in “A Simple Transmit Diversity Technique for Wireless Communications” IEEE Journal on Select Areas in Communications, vol. 16, no. 8, October 1998. Here, a case in which the number of transmitting antennas is two and the number of transmitted modulated signals is two is described by way of example, but the present invention is not limited to this case, and a space-time block coding method in which the number of transmitting antennas is increased is also shown in “Space-Time Block Codes from Orthogonal Design” IEEE Transactions on Information Theory, pp. 1456-1467, vol. 45, no. 5, July 1999, etc. Error correction coding such as convolutional coding is executed on each modulated signal. 
       FIG. 45 , in which parts corresponding to parts in  FIG. 4  are assigned the same codes as in  FIG. 4 , shows the configuration of the reception unit  4500  of a receiving apparatus of this embodiment. Signal separation section  4501  of reception unit  4500  has spread signal A channel fluctuation estimation signal  408  (corresponding to h 1  of Equation (9)), spread signal B channel fluctuation estimation signal  410  (corresponding to h 2  of Equation (9)), and despread received quadrature baseband signal  406  (corresponding to R 1 , R 2  of Equation (9)), as input, and by performing Equation (9) inverse matrix computation, finds baseband signal  4502  (baseband estimation signal corresponding to S 1  in Equation (9)) and baseband signal  4503  (baseband estimation signal corresponding to S 2  in Equation (9)), which it outputs. 
     An eigenvalue calculation section  4504  has spread signal A channel fluctuation estimation signal  408  and spread signal B channel fluctuation estimation signal  410  as input, creates an Equation (9) matrix using these, calculates an eigenvalue of that matrix, and outputs eigenvalue signal  4505 . 
     Soft decision calculation section  4506  has baseband signal  4502  and eigenvalue signal  4505  as input, and finds a soft decision value as shown in Equation (5) in the same way as in Embodiment 7. At this time, a soft decision value  4507  is found using a coefficient found from eigenvalue signal  4505 —for example, eigenvalue minimum power—for weighting coefficient Ca(t)×D(t) in Equation (5), and this soft decision value  4507  is output. Error correction section  4508  has soft decision value  4507  as input, executes error correction decoding processing on soft decision value  4507 , and outputs received digital signal  4509  obtained by this means. 
     Similarly, soft decision calculation section  4510  has baseband signal  4503  and eigenvalue signal  4505  as input, and finds a soft decision value as shown in Equation (6) in the same way as in Embodiment 7. At this time, a soft decision value  4511  is found using a coefficient found from eigenvalue signal  4505 —for example, eigenvalue minimum power—for weighting coefficient Cb(t)×D(t) in Equation (6), and this soft decision value  4511  is output. Error correction section  4512  has soft decision value  4511  as input, executes error correction decoding processing on soft decision value  4511 , and outputs received digital signal  4513  obtained by this means. 
     Thus, according to this embodiment, in a receiving apparatus that receives transmit signals combining convolutional code and space-time code, by weighting received signal soft decision values using space-time code channel fluctuation matrix eigenvalues, it is possible to give a soft decision value an appropriate likelihood based on effective reception power, enabling the error rate characteristics of decoded receive data to be improved. 
     That is to say, according to this embodiment, it has been shown that receive data error rate characteristics can also be improved in a case where convolutional coding and space-time block coding are combined, if soft decision values are weighted using eigenvalues in the same way as in Embodiment 7. 
     The method whereby soft decision values are weighted using channel fluctuation matrix eigenvalues according to the present invention is not limited to Embodiment 7 or this embodiment, but can be widely applied to cases where processing is performed that separates multiplexed modulated signals by means of computation using channel fluctuation matrices, convolutional coding or the like is further executed, and soft decision decoding is carried out. 
     Embodiment 18 
     In above-described Embodiment 5, it was proposed that a plurality of antenna received signal combinations be created, a channel fluctuation matrix be created for each combination, channel fluctuation matrix eigenvalues be calculated for each combination, and modulated signals be separated using the antenna received signals of each combination and the channel fluctuation matrix corresponding to that combination, and also that modulated signals separated in each combination be weighted and combined using the channel fluctuation estimation matrix eigenvalues used at the time of separation. 
     In contrast to this, while this embodiment is the same as Embodiment 5 in that a plurality of antenna received signal combinations are created, a channel fluctuation matrix is created for each combination, channel fluctuation matrix eigenvalues are calculated for each combination, and modulated signals are separated using the antenna received signals of each combination and the channel fluctuation matrix corresponding to that combination, this embodiment differs from Embodiment 5 in that the Euclidian distances (branch metric) between reception points of modulated signals separated in each combination and candidate points are weighted and combined using the channel fluctuation matrix eigenvalues used at the time of separation, and soft decision values after weighting and combining are determined. 
     In this embodiment, a case is described in which signals with the frame configurations shown in  FIG. 3 , transmitted from transmission unit  100  with the configuration shown in  FIG. 1 , are received. 
     The reception unit of this embodiment has the same configuration as reception unit  1800  in  FIG. 18  described in Embodiment 5, except for the configuration of signal processing section  1801  of reception unit  1800 . In this embodiment, therefore, only the configuration of the signal processing section will be described. 
       FIG. 46  shows the configuration of a signal processing section  4600  according to this embodiment. In the reception unit of this embodiment, signal processing section  1801  of reception unit  1800  in  FIG. 18  is replaced by signal processing section  4600  in  FIG. 46 . 
     Eigenvalue calculation section  4608  of signal processing section  4600  applies channel fluctuation estimation signals  408 ,  410 ,  418 , and  420  as a first group to an Equation (3) matrix, finds value P 1  with the smallest matrix eigenvalue power, and outputs this eigenvalue power P 1 . Similarly, eigenvalue calculation section  4608  applies channel fluctuation estimation signals  408 ,  410 ,  1408 , and  1410  as a second group to an Equation (3) matrix, finds value P 2  with the smallest matrix eigenvalue power, and outputs this eigenvalue power P 2 . Similarly, eigenvalue calculation section  4608  applies channel fluctuation estimation signals  418 ,  420 ,  1408 , and  1410  as a third group to an Equation (3) matrix, finds value P 3  with the smallest matrix eigenvalue power, and outputs this eigenvalue power P 3 . 
     A signal separation section  4601  applies signals  408 ,  410 ,  406 ,  418 ,  420 , and  416  to Equation (3) as a first group, and by performing this inverse matrix computation, finds spread signal A received quadrature baseband signal  4602  (Ra 1 ) and spread signal B received quadrature baseband signal  4605  (Rb 1 ), and outputs these signals  4602  and  4605 . Similarly, signal separation section  4601  applies signals  408 ,  410 ,  406 ,  1408 ,  1410 , and  1406  to Equation (3) as a second group, and by performing this inverse matrix computation, finds spread signal A received quadrature baseband signal  4603  (Ra 2 ) and spread signal B received quadrature baseband signal  4606  (Rb 2 ), and outputs these signals  4603  and  4606 . Similarly, signal separation section  4601  applies signals  418 ,  420 ,  416 ,  1408 ,  1410 , and  1406  to Equation (3) as a third group, and by performing this inverse matrix computation, finds spread signal A received quadrature baseband signal  4604  (Ra 3 ) and spread signal B received quadrature baseband signal  4607  (Rb 3 ), and outputs these signals  4604  and  4607 . 
     Soft decision value calculation section  4609  has spread signal A received quadrature baseband signal  4602  (Ra 1 ) and eigenvalue power signal (P 1 ) as input, finds soft decision value  4610  by weighting received quadrature baseband signal  4602  (Ra 1 ) with eigenvalue power signal (P 1 ), and outputs this soft decision value  4610 . The operation at this time will be described using  FIG. 47 . 
       FIG. 47  is a drawing showing the QPSK signal point arrangement in the in-phase I-orthogonal Q plane, in which reference numeral  4701  indicates QPSK signal points, and [0,0], [0,1], [1,0], and [1,1] indicate transmit bits. Reference numeral  4702  indicates the position of a received quadrature baseband signal, and here shows the position of spread signal A received quadrature baseband signal  4602  (Ra 1 ). The Euclidian distances between QPSK signal points  4701  and received quadrature baseband signal  4602  (Ra 1 ) are designated D 1 [0,0], D 1 [0,1], D 1 [1,0], and D 1 [1,1]. Soft decision value calculation section  4609  finds P 1 ×D 1   2  [0,0], P 1 ×D 1   2  [0,1], P 1 ×D 1   2  [1,0], and P 1 ×D 1   2  [1,1], and outputs these as soft decision value signal  4610 . 
     Similarly, soft decision value calculation section  4611  has spread signal A received quadrature baseband signal  4603  (Ra 2 ) and eigenvalue power signal (P 2 ) as input, finds soft decision value  4612  by weighting received quadrature baseband signal  4603  (Ra 2 ) with eigenvalue power signal (P 2 ), and outputs this soft decision value  4612 . Actually, if the Euclidian distances between QPSK signal points  4701  and received quadrature baseband signal  4603  (Ra 2 ) in  FIG. 47  are designated D 2 [0,0], D 2 [0,1], D 2 [1,0], and D 2 [1,1], soft decision value calculation section  4611  finds P 2 ×D 2   2  [0,0], P 2 ×D 2   2  [0,1], P 2 ×D 2   2  [1,0], and P 2 ×D 2   2  [1,1], and outputs these as soft decision value signal  4612 . 
     Similarly, soft decision value calculation section  4613  has spread signal A received quadrature baseband signal  4604  (Ra 3 ) and eigenvalue power signal (P 3 ) as input, finds soft decision value  4614  by weighting received quadrature baseband signal  4604  (Ra 3 ) with eigenvalue power signal (P 3 ), and outputs this soft decision value  4614 . Actually, if the Euclidian distances between QPSK signal points  4701  and received quadrature baseband signal  4604  (Ra 3 ) in  FIG. 47  are designated D 3 [0,0], D 3 [0,1], D 3 [1,0], and D 3 [1,1], soft decision value calculation section  4613  finds P 3 ×D 3   2  [0,0], P 3 ×D 3   2  [0,1], P 3 ×D 3   2  [1,0], and P 3 ×D 3   2  [1,1], and outputs these as soft decision value signal  4614 . 
     Thus, soft decision value calculation sections  4609 ,  4611 , and  4613  perform computations whereby the Euclidian distances between the reception points of modulated signals separated in each combination and candidate points are weighted using the channel fluctuation matrix eigenvalues used at the time of separation. 
     Decision section  4621  has soft decision value signals  4610 ,  4612 , and  4614  as input, and finds P 1 ×D 1   2  [0,0]+P 2 ×D 2   2  [0,0]+P 3 ×D 3   2  [0,0], P 1 ×D 1   2  [0,1]+P 2 ×D 2   2  [0,1]+P 3 ×D 3   2  [0,1], P 1 ×D 1   2  [1,0]+P 2 ×D 2   2  [1,0]+P 3 ×D 3   2  [1,0], and P 1 ×D 1   2  [1,1]+P 2 ×D 2   2  [1,1]+P 3 ×D 3   2  [1,1]. Then decision section  4621  searches for the smallest of the four values obtained, and, if, for example, P 1 ×D 1   2  [0,0]+P 2 ×D 2   2  [0,0]+P 3 ×D 3   2  [0,0] is the smallest value, decides that the transmit bits are [0,0], and outputs this as received digital signal  4622 . 
     The soft decision value calculations and decision operation for spread signal A by soft decision value calculation sections  4609 ,  4611 , and  4613  and decision section  4621  have been described above . For spread signal B, the same kind of soft decision value calculations and decision operation are performed by soft decision value calculation sections  4615 ,  4617 , and  4619 , and decision section  4623 , and received digital signal  4624  is obtained. 
     Thus, according to this embodiment, by creating a plurality of antenna received signal combinations, creating a channel fluctuation matrix for each combination, calculating channel fluctuation matrix eigenvalues for each combination, separating modulated signals using the antenna received signals of each combination and the channel fluctuation matrix corresponding to that combination, weighting and combining the Euclidian distances (branch metric) between reception points of modulated signals separated in each combination and candidate points using the channel fluctuation matrix eigenvalues used at the time of separation, and taking the candidate signal point for which the Euclidian distance is smallest as a reception point, bit decision processing can be performed in which likelihood can be made higher the greater the effective reception power of an antenna&#39;s received signals, and receive data error rate characteristics can be improved. 
     Thus, this embodiment coincides with Embodiment 5 in that antenna received signals are separated on a combination-by-combination basis, and separated antenna received signals are weighted and combined using eigenvalues on a combination-by-combination basis, but differs in the method of weighting and combining. 
     Comparing this embodiment with Embodiment 5, the method of Embodiment 5 has the advantage of having fewer computations to find Euclidian distances than this embodiment, with the result that the circuitry is smaller in scale. From the standpoint of error rate characteristics, on the other hand, this embodiment is superior to Embodiment 5. In any case, both this embodiment and Embodiment 5 enable excellent error rate characteristics to be obtained by using eigenvalues as weighting coefficients. 
     This embodiment can also be applied to OFDM communications. A case in which this embodiment is applied to OFDM communications can be considered as combining the descriptions of this embodiment and Embodiment 6. That is to say, the method of this embodiment should be performed on a subcarrier-by-subcarrier basis. 
     The method according to the present embodiment can be applied to cases where received signals are applied error correction codes such as convolutional code, turbo code, and low density parity code. The decoding in this case is executed by finding a branch metric and a path metric sequentially based on weighted results. 
     Embodiment 19 
     In this embodiment, a reception method is proposed in which error correction decoding processing is added to the reception method of Embodiment 18. That is to say, the transmitting side transmits signals subjected to error correction coding using convolutional code, etc., as described in Embodiment 7, and the receiving side weights and combines received signals using eigenvalues as described in Embodiment 18, and then performs error correction decoding processing. 
     A receiving apparatus of this embodiment has error correction coding sections  2001  and  2002  as shown in  FIG. 20  and described in Embodiment 7, and receives signals transmitted by transmission unit  2000  that transmits convolutional coded signals. 
     The reception unit of this embodiment has the same configuration as reception unit  1800  in  FIG. 18  described in Embodiment 5, except for the configuration of signal processing section  1801  of reception unit  1800 . In this embodiment, therefore, only the configuration of the signal processing section will be described. 
       FIG. 48  shows the configuration of a signal processing section  4800  according to this embodiment. In the reception unit of this embodiment, signal processing section  1801  of reception unit  1800  in  FIG. 18  is replaced by signal processing section  4800  in  FIG. 48 . 
     In signal processing section  4800  of this embodiment, decision sections  4621  and  4623  in  FIG. 46  described in Embodiment 18 are simply replaced by error correction sections  4801  and  4803 ; other parts are assigned the same codes as in  FIG. 46  and descriptions thereof are omitted. 
     Error correction section  4801  has soft decision value signals  4610 ,  4612 , and  4614  as input, finds a metric from P 1 ×D 1   2  [0,0]+P 2 ×D 2   2  [0,0]+P 3 ×D 3   2  [0,0], P 1 ×D 1   2  [0,1]+P 2 ×D 2   2  [0,1]+P 3 ×D 3   2  [0,1], P 1 ×D 1   2  [1,0]+P 2 ×D 2   2  [1,0]+P 3 ×D 3   2  [1,0], and P 1 ×D 1   2  [1,1]+P 2 ×D 2   2  [1,1]+P 3 ×D 3   2  [1,1], obtains a received digital signal  4802  by performing Viterbi decoding, for example, and performing error correction, and outputs this received digital signal  4802 . 
     In the same way as error correction section  4801 , error correction section  4803  also finds a metric from the Euclidian distances from candidate signal points weighted and combined by means of eigenvalues, obtains a received digital signal  4804  by performing error correction such as Viterbi decoding, for example, and outputs this received digital signal  4804 . 
       FIG. 49  shows simulation results for this embodiment. In this simulation, the relationship between Eb/No (bit-to-noise spectral density ratio) and BER (bit error rate) was investigated when using convolutional code and 2, 3, and 4 receiving antennas, as an example. In  FIG. 49 , reference numeral  4901  indicates the characteristic with two receiving antennas, reference numeral  4902  the characteristic with three receiving antennas, and reference numeral  4903  the characteristic with four receiving antennas. As can be seen from  FIG. 49 , using the configuration of this embodiment enables extremely good error rate characteristics to be obtained, especially in proportion to the number receiving antennas. 
     Thus, according to this embodiment, by performing error correction decoding processing in addition to providing the configuration of Embodiment 18, it is possible to obtain extremely good error rate characteristics. 
     In this embodiment, a method has been described that combines the method of Embodiment 18 with soft decision decoding, but the same kind of effect can also be obtained with a method combining the method of Embodiment 5 and soft decision decoding. 
     Other Embodiments 
     In the above-described embodiments, the descriptions have centered on a receiving apparatus that performs demodulation processing that takes effective reception power into consideration by using channel fluctuation matrix eigenvalues. Here, an eigenvalue may be used directly, or may be used after approximation. Approximation methods for finding an eigenvalue include a method whereby approximation is executed on channel fluctuation matrix elements, such as finding an eigenvalue by making the power of each element of a channel fluctuation matrix equal, for example. When approximation by making the power of each element of a channel fluctuation matrix equal is performed, an eigenvalue is found only at the phase of each element of a channel fluctuation matrix. Therefore, control of antenna selection, antenna combining, decoding, and so forth, is performed taking only the phase of each element of the channel fluctuation matrix into consideration. In this case, it is not necessarily essential to perform common control of the signal level of each antenna. 
     In other words, according to the decoding method using eigenvalues in the present embodiment, there are generally two methods of obtaining an eigenvalue that accurately reflects effective reception power. One method is to correct received signal levels so as to make the received signal levels at respective antennas virtually equal and to correct an eigenvalue in accordance with the received signal levels. The other method is to find an eigenvalue only from the phase of each element in channel fluctuation. 
     Also, in the above descriptions of a soft decision decoding method using eigenvalues, eigenvalue minimum power is used as a weighting coefficient, but the present invention is not limited to this, and it is also possible, for example, to input an eigenvalue and find a weighting coefficient from that eigenvalue. However, when eigenvalue minimum power is used as a weighting coefficient, receive data with extremely good error rate characteristics is obtained. 
     Furthermore, in the above-described embodiments, a case has been described in which soft decision decoding is performed with eigenvalue minimum power as a weighting coefficient, but error rate characteristics can also be improved if eigenvalue minimum power is used as a weighting coefficient in hard decision decoding. 
     The present invention is not limited to the above-described embodiments, and various variations and modifications may be possible without departing from the scope of the present invention. 
     This application is based on Japanese Patent Applications No. 2002-329453 filed on Nov. 13, 2002, No. 2002-374393 filed on Dec. 25, 2002, No. 2003-018761 filed on Jan. 28, 2003, and No. 2003-366249 filed on Oct. 27, 2003, entire content of which is expressly incorporated by reference herein.