Patent Publication Number: US-6710716-B1

Title: Power detecting circuit and demodulator comprising the same

Description:
TECHNICAL FIELD 
     The present invention relates to a power detector used in a communication apparatus for transmitting and receiving high frequency signals, or a measurement device for measuring signal levels of high frequency signals, and a demodulator using the same. 
     BACKGROUND ART 
     In a conventional high frequency power detector, a Schottky barrier diode has often been mainly used. 
     FIG. 1 is circuit diagram of an example of the configuration of a conventional high frequency power detector using the diode. 
     As shown in FIG. 1, this high frequency power detector  1  is comprised of a diode D 1  as an active element, a DC bias resistor R 1 , a capacitor C 1 , and a load resistor RL 1 . 
     An anode of the diode D 1  is connected to an input terminal Tin 1  of a high frequency signal RFin and one end of the resistor R 1 , while a cathode thereof is connected to an output terminal Tout 1 , one electrode of the capacitor C 1  for removing a high frequency component, and one end of the load resistor RL 1 . The other ends of the resistors R 1  and RL 1  and the other electrode of the capacitor Cl are grounded. 
     In the high frequency power detector  1  having such a configuration, the high frequency signal RFin is input to the input terminal Tin 1 . By a rectification function of the diode D 1  and the capacitor C 1  having a sufficiently large capacitance, an envelope component of the input high frequency signal is output as a detection output signal Vout. 
     In the high frequency power detector  1 , it is required to linearly obtain the detection output voltage Vout from a signal level as low as possible to a signal level as high as possible, that is in a wide dynamic range. 
     FIG. 2 is a diagram of an example of characteristics of the high frequency power detector using a diode as an active element. 
     This example plotted the relationship of the output voltage Vout with respect to an input high frequency power Pin obtained when a Schottky barrier diode was used, a bias voltage Vd of the diode D 1  in FIG. 1 was set at 0V (Vd=0V: zero bias), and the frequency of the high frequency signal was 10 GHz. 
     The conventional power detector using a Schottky barrier diode having such a characteristic has the following disadvantages. 
     In order to raise the detection performance, the circuit is produced by using a special semiconductor process. Accordingly, the conventional power detector is not suited for an integrated circuit. 
     For this reason, the conventional power detector has to have a hybrid configuration. This induces a rise of production costs, a restriction of the operation band, and an increase of production variability. 
     When the power detector is comprised by a semiconductor process enabling circuit integration, the detection characteristic thereof is deteriorated. 
     In recent years, there have been strong demands for reduction of size and lowering of price of mobile phones and other wireless communications devices. Circuit integration is important as a means for responding to such demands. 
     Therefore, in order to obtain a high performance, high frequency power detector suited for circuit integration, a power detector using a field effect transistor (FET) as an active element has been investigated (for example, the above document). 
     FIG. 3 is a circuit diagram of an example of the configuration of a conventional high frequency power detector using a silicon (Si) MOSFET. 
     As shown in FIG. 3, this high frequency power detector  2  is comprised of a field effect transistor (hereinafter, simply referred to as a “transistor”) Q 1 , resistors R 2  and R 3 , capacitors C 2  and C 3 , a voltage source V 1 , and a load resistor RL 2 . 
     In this high frequency power detector  2 , a gate of the transistor Q 1  is biased by a bias supply circuit comprised of the voltage source V 1 , resistor R 3 , and the capacitor C 2 . The input high frequency signal RFin is propagated through the transistor Q 1  having a predetermined resistance between the drain and the source, and an envelope component of the input high frequency signal is output as the detection output signal Vout by the capacitor C 3  having a large capacitance on the output side. 
     However, the high frequency power detector of FIG. 3 has the following disadvantages. 
     Since it uses an SiMOSFET, the maximum operation frequency is low, i.e., the 1.5 GHz band. 
     Also, as shown in FIG. 4, there is room for improvement of linearity of the input power versus detection output voltage characteristic (Mohamed RATNI, Bernard HUYART, et al., “RF Power Detector using a Silicon MOSFET”, International Microwave Symposium, 1998). 
     Also, in the power detector  2 , where the output format is the single end system and the latter stage of the linear detector has a balance input, an additional unbalance/balance conversion circuit becomes necessary. 
     FIG. 5 is a circuit diagram of another example of the configuration of the high frequency power detector using a field effect transistor as an active element (refer to Japanese Unexamined Patent Publication (Kokai) No. 10-234474). 
     As shown in FIG. 5, this high frequency power detector  3  is comprised by a transistor (FET) Q 2 , a DC cutting capacitor Cin, a bias resistor R 4 , voltage sources V 2  and V 3 , a load resistor RL 3 , an output side capacitor C 4 , a coupling capacitor Cd, and an inductor Ld. A gate bias supply circuit  3   a  is comprised by the resistor R 4 , while a drain bias supply circuit  3   b  is comprised by the inductor Ld. 
     In this high frequency power detector  3 , the high frequency signal RFin input to an input terminal Tin 3  is supplied via the DC cutting capacitor Cin to the gate of the transistor Q 2 . The gate of the transistor Q 2  is supplied with the gate bias voltage of the gate bias supply circuit  3   a  connected to the voltage source V 2  for supplying a voltage Vgg. Also, the drain of the transistor Q 2  has connected to it the drain bias supply circuit  3   b  for supplying the drain bias voltage. Note that the voltage source V 3  for supplying a DC voltage Vdd is connected to the drain bias supply circuit  3   b.    
     A coupling capacitor Cd having a sufficiently large capacitance value is connected between the drain of the transistor Q 2  and a ground potential GND. The resistor RL 3  and the coupling capacitor C 4  having a sufficiently large capacitance value are connected in parallel between the source of the transistor Q 2  and the ground potential GND. Then, a potential difference Vout between the transistor Q 2  and the ground potential GND becomes the detection output signal. 
     FIG. 6 shows the detection characteristics of the high frequency power detector of FIG.  5 . 
     This power detector  3  enables the realization of a detector of a small size and low cost and adapted to broadband high frequency operation, but has the following disadvantages. 
     As shown in FIG. 6, the fluctuation of the detection output voltage versus input power characteristic is large compared with the gate-source bias fluctuation. 
     As shown in FIG. 6, depending on the bias conditions, sometimes a DC offset occurs. 
     When a pinchoff voltage of the transistor Q 2  fluctuates due to a production variability, temperature fluctuation, etc., the fluctuation of the detection output voltage versus input voltage characteristic is large. 
     Also, in the power detector  3 , when the output format is the single end system and the latter stage of the linear detector has a balance input, an additional unbalance/balance conversion circuit becomes necessary. 
     DISCLOSURE OF THE INVENTION 
     The present invention was made in consideration of such a circumstance and has as an object thereof to provide a high performance power detector not only suited for monolithic structures, small in size, low in cost, and suited for broadband high frequency operation, but also excellent in the linearity of the detection characteristic relative to the bias fluctuation, having a small fluctuation of the detection characteristic relative to the FET threshold voltage fluctuation, having a small DC offset, and not requiring an a additional circuit even when the latter stage circuit has a balance input, and a demodulator using the same. 
     A first aspect of the present invention is a power detector for detecting a signal level of a high frequency signal, having a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to the gate of the second field effect transistor, a resistor connected between a connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential, a drain bias supply circuit for supplying the drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein a voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     A second aspect of the present invention is a power detector for detecting a signal level of a high frequency signal, having a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to a gate of the second field effect transistor, a third field effect transistor connected between the connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential, a third gate bias supply circuit for supplying the gate bias voltage to the gate of the third field effect transistor, a drain bias supply circuit for supplying the drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     A third aspect of the present invention is a power detector for detecting a signal level of a high frequency signal, having a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to the gate of the second field effect transistor, a first resistor and a second resistor connected in series between the connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential and having the related connecting point supplied with the high frequency signal, an inductor connected between the connecting point of the first resistor and second resistor and a reference potential, a drain bias supply circuit for supplying the drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and the reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     Preferably, the first field effect transistor and second field effect transistor have substantially identical characteristics, the drain bias supply circuit includes a first drain bias resistor connected between the drain of the first field effect transistor and the voltage source and a second drain bias resistor connected between the drain of the second field effect transistor and the voltage source, a resistance value of the first drain bias resistor and the resistance value of the second drain bias resistor are set at substantially equal values, and a capacitance value of the first capacitor and the capacitance value of the second capacitor are set. at substantially equal values. 
     Also, preferably, a ratio Wga/Wgb of a gate width Wga of the first field effect transistor and a gate width Wgb of the second field effect transistor is set to N, the drain bias supply circuit includes the first drain bias resistor connected between the drain of the first field effect transistor and the voltage source and the second drain bias resistor connected between the drain of the second field effect transistor and the voltage source, the first gate bias voltage of the first gate bias supply circuit and the second gate bias voltage of the second gate bias supply circuit are set to substantially equal, a resistance value Ra of the first drain bias resistor and a resistance value Rb of the second drain bias resistor are set so as to satisfy a condition of Ra/Rb=1/N, and the capacitance value of the first capacitor and the capacitance value of the second capacitor are set to substantially equal values. 
     A fourth aspect of the present invention is a power detector for detecting a signal level of a high frequency signal, having a first field effect transistor having a gate supplied with the high frequency signal and a source connected to a reference potential, a second field effect transistor having a source connected to a reference potential, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to a gate of the second field effect transistor, a drain bias supply circuit for supplying a drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     Preferably, the first field effect transistor and second field effect transistor have substantially identical characteristics, the drain bias supply circuit includes a first drain bias resistor connected between the drain of the first field effect transistor and the voltage source and a second drain bias resistor connected between the drain of the second field effect transistor and the voltage source, the first gate bias voltage of the first gate bias supply circuit and the gate bias voltage of the second gate bias supply circuit are substantially equal and set to a voltage substantially equal to the threshold voltage of the first and second field effect transistors, the resistance value of the first drain bias resistor and the resistance value of the second drain bias resistor are set at substantially equal values, and the capacitance value of the first capacitor and the capacitance value of the second capacitor are set at substantially equal values. 
     Also, preferably, the ratio Wga/Wgb of the gate width Wga of the first field effect transistor and the gate width Wgb of the second field effect transistor is set to N, the drain bias supply circuit includes the first drain bias resistor connected between the drain of the first field effect transistor and the voltage source and the second drain bias resistor connected between the drain of the second field effect transistor and the voltage source, the first gate bias voltage of the first gate bias supply circuit and the second gate bias voltage of the second gate bias supply circuit are set to voltages which are substantially equal to each other and substantially equal to the threshold voltage of the first and second field effect transistors, the resistance value Ra of the first drain bias resistor and the resistance value Rb of the second drain bias resistor are set so as to satisfy the condition of Ra/Rb=1/N, and the capacitance value of the first capacitor and the capacitance value of the second capacitor are set to substantially equal values. 
     Also, a demodulator according to a fifth aspect of the present invention has a first signal input terminal to which a first high frequency signal is input, a second signal input terminal to which a second high frequency signal is input, a generating means for generating two high frequency signals having a phase difference based on at least one high frequency signal between the first high frequency signal input from the first signal input terminal and the second high frequency signal input from the second signal input terminal and including at least one output terminal for outputting generated high frequency signals, at least one power detector for receiving as the input the high frequency signals output from the output terminal of the generating means and detecting the signal level of the input high frequency signals, and a conversion circuit for converting the output signal of the power detector to a plurality of signal components contained in the first or second high frequency signal, wherein the power detector has a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to a gate of the second field effect transistor, a resistor connected between the connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential, a drain bias supply circuit for supplying a drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     A demodulator according to a sixth aspect of the present invention has a first signal input terminal to which a first high frequency signal is input, a second signal input terminal to which a second high frequency signal is input, a generating means for generating two high frequency signals having a phase difference based on at least one high frequency signal between the first high frequency signal input from the first signal input terminal and the second high frequency signal input from the second signal input terminal and including at least one output terminal for outputting the generated high frequency signals, at least one power detector for receiving as input the high frequency signals output from the output terminal of the generating means and detecting the signal level of the input high frequency signals, and a conversion circuit for converting the output signal of the power detector to a plurality of signal components contained in the first or second high frequency signal, wherein the power detector has a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to the gate of the second field effect transistor, a third field effect transistor connected between the connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential, a third gate bias supply circuit for supplying a gate bias voltage to a gate of the third field effect transistor, a drain bias supply circuit for supplying a drain bias voltage to drains of the first field effect transistor and second electric field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     A demodulator according to a seventh aspect of the present invention has a first signal input terminal to which a first high frequency signal is input, a second signal input terminal to which a second high frequency signal is input, a generating means for generating two high frequency signals having a phase difference based on at least one high frequency signal between the first high frequency signal input from the first signal input terminal and the second high frequency signal input from the second signal input terminal and including at least one output terminal for outputting generated high frequency signals, at least one power detector for receiving as input the high frequency signals output from the output terminal of the generating means and detecting the signal level of the input high frequency signals, and a conversion circuit for converting the output signal of the power detector to a plurality of signal components contained in the first or second high frequency signal, wherein the power detector has a first field effect transistor having a gate supplied with the high frequency signal, a second field effect transistor having a source connected to a source of the first field effect transistor, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to a gate of the second field effect transistor, a first resistor and a second resistor connected in series between the connecting point of sources of the first field effect transistor and second field effect transistor and a reference potential and having the related connecting point supplied with the high frequency signal, an inductor connected between the connecting point of the first resistor and second resistor and the reference potential, a drain bias supply circuit for supplying a drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     A demodulator according to an eighth aspect of the present invention has a first signal input terminal to which a first high frequency signal is input, a second signal input terminal to which a second high frequency signal is input, a generating means for generating two high frequency signals having a phase difference based on at least one high frequency signal between the first high frequency signal input from the first signal input terminal and the second high frequency signal input from the second signal input terminal and including at least one output terminal for outputting generated high frequency signals, at least one power detector for receiving as input the high frequency signals output from the output terminal of the generating means and detecting the signal level of the input high frequency signals, and a conversion circuit for converting the output signal of the power detector to a plurality of signal components contained in the first or second high frequency signal, wherein the power detector has a first field effect transistor having a gate supplied with the high frequency signal and a source connected to a reference potential, a second field effect transistor having a source connected to a reference potential, a first gate bias supply circuit for supplying a gate bias voltage to the gate of the first field effect transistor, a second gate bias supply circuit for supplying a gate bias voltage to a gate of the second field effect transistor, a drain bias supply circuit for supplying a drain bias voltage to drains of the first field effect transistor and second field effect transistor, a first capacitor connected between the drain of the first field effect transistor and a reference potential, and a second capacitor connected between the drain of the second field effect transistor and a reference potential, wherein the voltage difference between the drain voltage of the first field effect transistor and the drain voltage of the second field effect transistor is defined as the detection output. 
     According to the present invention, in the power detector, the first field effect transistor and the second field effect transistor are used as active elements. 
     The high frequency signal is supplied via for example a matching circuit or DC (direct current) cutting capacitor to the gate of the first field effect transistor. 
     Also, the gate of the first field effect transistor is supplied with a gate bias voltage of the first gate bias supply circuit. Similarly, the gate of the second field effect transistor is supplied with for example a gate bias voltage substantially equal to the first gate bias voltage of the second gate bias supply circuit. 
     Also, the drains of the first field effect transistor and second field effect transistor are supplied with drain bias voltages via for example resistors having substantially equal resistance values. 
     Between the drains of the first field effect transistor and the second field effect transistor and the reference potential (ground potential), first and second capacitors having sufficiently large capacitance values are connected, so the drains of the first field effect transistor and second field effect transistor exhibit a stable state in terms of high frequency, and the voltage difference between the voltage of the drain of the first field effect transistor and the voltage of the drain of the second field effect transistor is supplied as the detection output signal to for example the conversion circuit of the latter stage. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram of an example of the configuration of a conventional high frequency power detector using a diode. 
     FIG. 2 is a diagram of an example of characteristics of a high frequency power detector using a diode as an active element. 
     FIG. 3 is a circuit diagram of an example of the configuration of a conventional high frequency power detector using a silicon (Si) MOSFET. 
     FIG. 4 is a diagram of an example of characteristics of the high frequency power detector of FIG. 3 using an SiMOSFET. 
     FIG. 5 is a circuit diagram of another example of the configuration of a high frequency power detector using a field effect transistor as an active element. 
     FIG. 6 is a diagram of a detection characteristic of a high frequency power detector using a field effect transistor as the active element of FIG.  5 . 
     FIG. 7 is a circuit diagram of a first embodiment of a high frequency power detector according to the present invention. 
     FIG. 8 is a circuit diagram of a second embodiment of a high frequency power detector according to the present invention. 
     FIG. 9 is a circuit diagram of a third embodiment of a high frequency power detector according to the present invention. 
     FIG. 10 is a circuit diagram of a fourth embodiment of a high frequency power detector according to the present invention. 
     FIG. 11 is a circuit diagram of a fifth embodiment of a high frequency power detector according to the present invention. 
     FIG. 12 is a diagram of an example of detection characteristics of the high frequency power detector of FIG.  11 . 
     FIG. 13 is a diagram of a high frequency input power Pin versus output detection voltage Vout characteristic when gate bias voltages VggA and VggB are used as parameters in the circuit of FIG.  11  and corresponds to FIG. 6 showing the characteristics of the conventional example. 
     FIG. 14 is a circuit diagram of a sixth embodiment of a high frequency power detector according to the present invention. 
     FIG. 15 is a circuit diagram of a seventh embodiment of a high frequency power detector according to the present invention. 
     FIG. 16 is a circuit diagram of an example of the configuration of a 3-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     FIG. 17 is a circuit diagram of an example of the configuration of a 4-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     FIG. 18 is a circuit diagram of an example of the configuration of a 5-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     FIG. 19 is a circuit diagram of an example of the configuration of a 6-port demodulator to which the high frequency power detector according to the present invention can be applied. 
    
    
     BEST MODE FOR WORKING THE INVENTION 
     Below, an explanation will be made of embodiments of the present invention by referring to the attached drawings. 
     FIG. 7 is a circuit diagram of a first embodiment of a high frequency power detector according to the present invention. 
     A high frequency power detector  100  according to the present first embodiment is comprised of two first and second field effect transistors (hereinafter, referred to as “transistors”) Q 101  and Q 102  as the active elements, capacitors C 101 , C 102 , and C 103 , resistors R 101 , R 102 , R 103 , R 104 , and R 105 , voltage sources V 101 , V 102 , and V 103 , and a matching circuit (MTR)  101 . 
     The gate of the transistor Q 101  is connected to one electrode of the DC cutting capacitor C 101 , and the other electrode of the capacitor C 101  is connected via the matching circuit  101  to an input terminal TIN 101  of the high frequency signal RFin. 
     One end of the resistor R 101  is connected to the gate of the transistor Q 101 , while the other end of the resistor R 101  is connected to the voltage source V 101  of the voltage VggA. A first gate bias supply circuit  102  for supplying the gate bias voltage of the transistor Q 101  is comprised by this resistor R 101 . 
     One end of the resistor  102  is connected to the gate of the transistor Q 102 , while the other end of the resistor is connected to the voltage source V 102  of the voltage VggB. A second gate bias supply circuit  103  for supplying the gate bias voltage of the transistor Q 102  is comprised by this resistor R 102 . 
     The source of the transistor Q 101  and the source of the transistor Q 102  are connected. The connecting point thereof is connected via the resistor R 103  to the ground potential GND. 
     The drain of the transistor Q 101  is connected to one end of the resistor R 104 , one electrode of the capacitor C 102 , and a first output terminal TOT 101 . The other end of the resistor R 104  is connected to the voltage source V 103  of the voltage Vdd, while the other electrode of the capacitor C 102  is connected to the ground potential GND. 
     The drain of the transistor Q 102  is connected to one end of the resistor R 105 , one electrode of the capacitor C 103 , and a second output terminal TOT 102 . The other end of the resistor R 105  is connected to the voltage source V 103  of the voltage Vdd, while the other electrode of the capacitor C 103  is connected to the ground potential GND. 
     The drain bias voltage is supplied to the drain of the transistor Q 101  via the resistor R 104 , while the drain bias voltage is supplied to the drain of the transistor Q 102  via the resistor R 105 . 
     In the high frequency power detector  100  comprised by such a connection configuration, the transistors Q 101  and Q 102  serving as the active elements have an identical device structure so that they have almost the same characteristics. 
     The gate bias voltage of the gate bias supply circuit  102  and the gate bias voltage of the gate bias supply circuit  103  are set to substantially equal values. For example, the DC voltage VggA of the voltage source V 101  and the DC voltage VggB of the voltage source V 102  are set to substantially equal values, while the resistance value of the resistor R 101  and the resistance value of the resistor R 102  are set to substantially equal values. 
     Also, a resistance value Rda of the resistor R 104  connected to the drain of the transistors Q 101  and a resistance value Rdb of the resistor R 105  connected to the drain of the transistor Q 102  satisfy the condition of Rda=Rdb. Similarly, a capacitance value Couta of the capacitor C 102  and a capacitance value Coutb of the capacitor C 103  satisfy a condition of Couta=Coutb. 
     Alternatively, in the high frequency power detector  100 , when the ratio of the gate width Wga of the transistor Q 101  and the gate width Wgb of the transistor Q 102  (Wga/Wgb) is N, conditions of Rda/Rdb=1/N and Couta=Coutb are satisfied. 
     The capacitance values Couta and Coutb of the capacitors C 102  and C 103  are set at values large enough to give impedances of almost 0 ohm at higher frequencies including the input high frequency signal RFin of a frequency fin. 
     Also, in the gate bias supply circuits  102  and  103 , desirably isolation is established between the gates of the transistors Q 101  and Q 102  and the voltage source at the input signal frequency fin. 
     Next, an explanation will be made of the operation by the above configuration. 
     The high frequency signal RFin input to the input terminal TIN 101  is supplied via the matching circuit  101  and the DC (direct current) cutting capacitor C 101  to the gate of the transistor Q 101 . 
     Also, the gate of the transistor Q 101  is supplied with the gate bias voltage of the gate bias supply circuit  102  connected to the voltage source V 101  for supplying the voltage Vgg. Similarly, the gate of the transistor Q 102  is supplied with the gate bias voltage of the gate bias supply circuit  103  connected to the voltage source V 102  for supplying the voltage Vgg. 
     Also, the drains of the transistors Q 101  and Q 102  are supplied with the drain bias voltages via the resistors R 104  and R 105 . 
     Between the drains of the transistors Q 101  and Q 102  and the ground potential GND, coupling capacitors C 102  and C 103  having sufficiently large capacitance values are connected, so the drains of the transistors Q 101  and Q 102  exhibit a stable state in terms of the high frequency, and the voltage difference between the voltage of the drain of the transistor Q 101 , that is, the voltage of the first output terminal TOT 101 , and the voltage of the drain of the transistor Q 102 , that is, the second output terminal TOT 102 , is supplied as the detection output signal to a not illustrated processing circuit of a latter stage. 
     As explained above, according to the first embodiment, the configuration was made so that two transistors (FET) Q 101  and Q 102  having substantially the same characteristics with sources connected to each other and connecting the resistor R 103  as the current source to their connecting point were used as the active elements, substantially equal gate bias voltages were supplied to the gates of the transistors Q 101  and Q 102  by the gate bias supply circuits  102  and  103 , and then substantially equal drain bias voltages were supplied to the drains of the transistors Q 101  and Q 102 , and further capacitors C 102  and C 103  having capacitance values set at values which were substantially equal and large enough to give impedances of almost 0 ohm at higher frequencies including the input high frequency signal RFin were connected between the drains of the transistors Q 101  and Q 102  and the ground, the high frequency signal RFin was supplied to the gate of the transistor Q 101 , and the voltage difference between the drain of the transistor Q 101  and the drain of the transistor Q 102  was used as the detection output, so there are the following effects. 
     Namely, in comparison with a conventional detector using a Schottky barrier diode, the circuit can be comprised on a semiconductor process suited for high frequency for example, GaAs, so is suited for monolithic structures. Accordingly, a detector of a small size, low cost, and suitability for broadband high frequency operation can be realized. 
     Also, the power detector of FIG. 7 has the advantage of a high performance, high frequency detector, excellent in the linearity of the detection characteristic when compared with a conventional power detector, having a smaller fluctuation relative to the bias fluctuation, having a smaller fluctuation of detection characteristic relative to the FET threshold voltage fluctuation, and having a small DC offset. 
     Also, since the power detector of FIG. 7 has a balance output, there is an advantage that the connection becomes simple when the latter stage circuit has a balance input. 
     FIG. 8 is a circuit diagram of a second embodiment of a high frequency power detector according to the present invention. 
     The difference of the present second embodiment from the first embodiment resides in that a transistor Q 103  serving as a third FET having a gate with the bias voltage supplied thereto by a third gate bias supply circuit  104  is connected between the connecting point of sources of the transistors Q 101  and Q 102  and the ground terminal GND in place of connecting the resistor. 
     The gate bias supply circuit  104  is comprised of a resistor R 106  connected between the gate of the transistor Q 104  and a voltage source V 104  of a voltage VggC. 
     The rest of the configuration of the second embodiment is similar to the first embodiment. 
     According to the second embodiment, similar effects to those of the first embodiment can be obtained. 
     FIG. 9 is a circuit diagram of a third embodiment of a high frequency power detector according to the present invention. 
     The difference of the third embodiment from the first embodiment resides in that the input format employs not an unbalance input, but a balance input. 
     For this reason, in a high frequency power detector  100 B according to the third embodiment, two input terminals TIN 101  and TIN 102  are connected to the input side of a matching circuit  101   a,  and two DC cutting capacitors C 101   a  and C 101   b  are connected to the output side. Further, between the other end of the resistor R 103  having one end connected to the connecting point of the sources of the transistors Q 101  and Q 102  and the ground potential GND, a resistor R 107  and an inductor L 101  are connected in parallel. A high frequency signal via the DC cutting capacitor C 101   b  is supplied to the connecting point of the other end of the resistor R 103 , resistor R 107 , and the inductor L 101 . The reason for the connection of the resistor R 107  to the other end side (ground side) of the resistor R 103  is that the signal can be extracted in a state where it floats in terms of high frequency. Also, the inductor L 101  was connected in parallel to the resistor R 107  for achieving a short circuit state in terms of DC (direct current) since the current changes by the resistor R 107  when the signal is input and deterioration of sensitivity is induced. 
     The rest of the configuration of the present third embodiment is similar to the first embodiment. 
     According to the third embodiment, similar effects to those of the first embodiment can be obtained. 
     FIG. 10 is a circuit diagram of a fourth embodiment of a high frequency power detector according to the present invention. 
     The difference of the fourth embodiment from the first embodiment resides in that the sources of the transistors Q 101  and Q 102  serving as the active elements are directly connected to the ground potential GND in place of the connection to the ground potential GND via the resistors. 
     In this case, the gate bias voltages supplied to the gates of the transistors Q 101  and Q 102  are set at values in the vicinity of the threshold voltage of the transistors Q 101  and Q 102 . 
     Concretely, the transistors Q 101  and Q 102  have an identical device structure and gate bias voltages VggA=VggB and VggB≈Vth (Vth is the threshold voltage of the transistors Q 101  and Q 102 ). 
     Also, the resistance value Rda of the resistor R 104  and the resistance value Rdb of the resistor R 105  connected to the drains of the transistors Q 101  and Q 102  satisfy the condition of Rda=Rdb. Similarly, the capacitance value Couta of the capacitor C 102  and the capacitance value Coutb of the capacitor C 103  satisfy the condition of Couta=Coutb. 
     Alternatively, in the high frequency power detector  100 C, when the ratio of the gate width Wga of the transistor Q 101  and the gate width Wgb of the transistor Q 102  (Wga/Wgb) is N, conditions of VggA=VggB≈Vth, Rda/Rdb=1/N, and Couta=Coutb are satisfied. 
     The rest of the configuration of the fourth embodiment is similar to the first embodiment. 
     According to the fourth embodiment, there are the advantages that, not only can effects similar to those by the first embodiment be obtained, but also a reduction of power consumption can be achieved. 
     FIG. 11 is a circuit diagram of a fifth embodiment of a high frequency power detector according to the present invention. 
     The difference of the fifth embodiment from the first embodiment resides in that the gate bias supply circuits  102 A and  103 A for the transistors Q 101  and Q 102  are comprised so as to generate bias voltages by dividing the resistance of the voltage Vdd of the drain bias voltage source V 103 . 
     The first gate bias supply circuit  102 A is comprised of a resistor R 101   a  and a resistor R 101   b  connected in series between the voltage source V 103  and the ground potential GND. The connecting point of the resistors R 101   a  and R 101   b  is connected to one electrode of the DC cutting capacitor C 101  and the gate of the transistor Q 101 . 
     A second gate bias supply circuit  103 A is comprised of a resistor R 102   a  and a resistor R 102   b  connected in series between the voltage source V 103  and the ground potential GND. The connecting point of the resistors R 102   a  and R 102   b  is connected to the gate of the transistor Q 102 . 
     Also, in the fifth embodiment, as the matching circuit  101 , an example of a matching circuit comprised by two concentrated constant elements is shown. 
     In the example of FIG. 11, as a concentrated constant element, use is made of an inductor L 102  connected between the input terminal TIN 101  and the other electrode of the DC cutting capacitor C 101  and of a capacitor C 104  connected between the connecting point of the inductor L 102  and the capacitor C 101  and the ground potential GND. 
     Note that, according to the object of use of the detector, there also exists a case where resistor matching and a matching circuit comprised by a distributed constant element or the like become necessary. 
     Also in the high frequency power detector  100 D, the high frequency signal RFin of the frequency fin input to the input terminal TIN 101  is supplied via the matching circuit  101  and the DC cutting capacitor C 101  to the gate of the transistor Q 101 . 
     At this time, the gate bias voltages generated at the gate bias supply circuits  102   a  and  103   a  by the resistor division are supplied to the gates of the transistors Q 101  and Q 102 . 
     In this way, the gate bias supply circuits  102   a  and  103   a  generate bias voltages by resistance division, so it is not necessary to establish isolation between the gates of the transistors Q 101  and Q 102  and the voltage source at the input signal frequency fin as in the case of the first embodiment. 
     Note that, it is also possible to comprise the gate bias supply circuit not only by resistance division, but by, for example, a choke coil (inductor having a sufficiently large inductance value), choke coil and shunt coupled capacitance, or distributed constant line, etc. Also, it becomes necessary for the circuit according to the fifth embodiment to achieve the same characteristics of the transistors Q 101  and Q 102 . 
     Also, in the circuit according to the fifth embodiment, the resistance values Rga 1  and Rgb 1  of the resistors R 101   a  and R 101   b  comprising the gate bias supply circuits  102   a  and  103   a  and resistance values Rga 2  and Rgb 2  of the resistors R 102   a  and R 102   b  must satisfy conditions of Rga 1 =Rga 2  and Rgb 1 =Rgb 2 , and the gate bias voltages of the transistors Q 101  and Q 102  must be made as equal as possible. 
     Also, the resistance value Rda of the resistor R 104  and the resistance value Rdb of the resistor R 105  connected to the drains of the transistors Q 101  and Q 102  satisfy the condition of Rda=Rdb. Similarly, desirably the capacitance value Couta of the capacitor C 102  and the capacitance value Coutb of the capacitor C 103  satisfy the condition of Couta=Coutb, and the capacitance values Couta and Coutb are set to capacitance values large enough to give impedances of almost 0 ohm at higher frequencies including the input high frequency signal of the input frequency fin. 
     Then, the voltage difference between the drain of the transistor Q 101  and the drain of the transistor Q 102  becomes the detection output Vout. 
     Below, the detection characteristics of the high frequency power detector of FIG. 11 will be considered in connection with the drawings. 
     FIG. 12 is a diagram of an example of the detection characteristics of the high frequency power detector of FIG.  11 . 
     In FIG. 12, the abscissa represents the input high frequency power Pin, and the ordinate represents the output detection voltage Vout. The frequency of the input high frequency signal is 5.5 GHz. Also, in FIG. 12, the characteristic of the power detector of FIG. 11 is indicated by a curve &lt; 1 &gt;, and the characteristic of the power detector using a FET of FIG. 5 is indicated by a curve &lt; 2 &gt; as comparative data. 
     As seen from FIG. 12, the power detector of FIG. 11 has a good linearity in comparison with the power detector of FIG.  5 . 
     Also, FIG. 13 is a diagram of a high frequency input power Pin versus output detection voltage Vout characteristic when the gate bias voltages VggA and VggB are used as parameters and corresponds to FIG. 6 showing characteristics of the conventional example. 
     Also in FIG. 13, the abscissa represents the input high frequency power Pin, while the ordinate represents the output detection voltage Vout. 
     It is learned that in the characteristic of the power detector according to the fifth embodiment shown in FIG. 13, compared with the characteristic of the conventional power detector shown in FIG. 5, the fluctuation of the Pin versus Vout characteristic is smaller relative to the gate bias fluctuation. 
     Concretely, in FIG. 6 showing the characteristics of the conventional circuit, when Vgs is large, for example when Vgs=±0.1V or more, a DC offset voltage is generated when Pin is small. 
     Contrary to this, in FIG. 13 showing the characteristics of the power detector according to the fifth embodiment, the DC offset is not generated. 
     Accordingly, the results of FIG. 13 show that, in the power detector according to the fifth embodiment, the variability of the Pin versus Vout characteristic is small in comparison with the conventional power detector such as FIG. 5 even if the threshold voltage of the FET varies. 
     According to the fifth embodiment, similar to the first embodiment, in comparison with the conventional power detector using a silicon Schottky diode, the circuit can be comprised on a semiconductor process suited for high frequency such as GaAs, so it is suitable for monolithic structures. Accordingly, a power detector of a small size, low cost, and suitability for a broadband high frequency operation can be realized. 
     Also, the power detector of FIG. 11 has the advantage that a high performance power detector excellent in the linearity of the detection characteristic, having a fluctuation of detection characteristic smaller relative to the bias fluctuation, having a small fluctuation of detection characteristic relative to the FET threshold voltage fluctuation, and in addition having a small DC offset in comparison with the conventional detector can be realized. 
     FIG. 14 is a circuit diagram of a sixth embodiment of a high frequency power detector according to the present invention. 
     The difference of the sixth embodiment from the fifth embodiment resides in that the consumed current is enhanced by setting the gate width Wgb of the transistor Q 102   a  smaller than the gate width Wga of the transistor Q 101   a,  and then setting the resistance value Rdb of the drain bias resistor R 105  larger than the resistance value Rda of the resistor R 104 . 
     Note that the power detector  1 OOE shown in FIG. 14 is a similar circuit as an equivalent circuit to the power detector  100 D shown in FIG.  11 . 
     In the power detector of FIG. 11 according to the fifth embodiment, it is necessary to make the characteristics of the transistors Q 101  and Q 102  equal. Therefore, the gate width Wga of the transistor Q 101  and the gate width Wgb of the transistor Q 102  are equal. 
     Contrary to this, in the power detector  100 E of FIG. 14 according to the present sixth embodiment, the ratio Wga/Wgb of the gate width Wga of the transistor Q 101  and the gate width Wgb of the transistor Q 102  is set at N, and further the resistance value Rdb of the resistor R 105   a  is set at N times the resistance value Rda of the resistor R 104   a.    
     By this, in the power detector  100 E of FIG. 14 according to the sixth embodiment, in comparison with the power detector  100 D of FIG. 10, the current consumption can be reduced to (N+1)/(2N) times. 
     Namely, according to the sixth embodiment, there are advantages in that not only can effects similar to those in the fifth embodiment be obtained, but also the current consumption can be reduced. 
     FIG. 15 is a circuit diagram of a seventh embodiment of a high frequency power detector according to the present invention. 
     The differences of the seventh embodiment from the second embodiment reside in that, similar to the fifth embodiment, the gate bias supply circuits  102   a  and  103   a  for the transistors Q 101  and Q 102  are comprised so as to generate bias voltages by the resistance division of the voltage Vdd of the drain bias voltage source V 103  and in that the third gate bias supply circuit  104   a  for the transistor Q 103  serving as the current source is comprised so as to generate bias voltage by the resistance division of the voltage Vdd of the drain bias voltage source V 103 . 
     The first gate bias supply circuit  102   a  is comprised of the resistor R 101   a  and the resistor R 101   b  connected in series between the voltage source V 103  and the ground potential GND. The connecting point of the resistors R 101   a  and R 101   b  is connected to one electrode of the DC cutting capacitor C 101  and the gate of the transistor Q 101 . 
     The second gate bias supply circuit  103   a  is comprised by the resistor R 102   a  and the resistor R 102   b  connected in series between the voltage source V 103  and the ground potential GND. The connecting point of the resistors R 102   a  and R 102   b  is connected to the gate of the transistor Q 102 . 
     The third gate bias supply circuit  104   a  is comprised by the resistor R 106   a  and the resistor R 106   b  connected in series between the voltage source V 103  and the ground potential GND. The connecting point of the resistors R 106   a  and R 106   b  is connected to the gate of the transistor Q 103 . 
     The rest of the configuration of the seventh embodiment is similar to that of the second embodiment. 
     According to the seventh embodiment, similar effects to those by the first and fifth embodiments can be obtained. 
     Various aspects of the high frequency power detector according to the present invention were explained above as the first to seventh embodiments. 
     Below, an explanation will be made of an N-port demodulator to which these high frequency power detector circuits according to the present invention can be applied. Note that, in the following explanation, in place of the seven reference numerals of  100  and  100 A to  100 F used in the above description, the high frequency power detector (PD) is indicated by the reference numeral  100 G including all circuits. 
     FIG. 16 is a circuit diagram of an example of the configuration of a 3-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     This 3-port demodulator  200  is comprised by using one high frequency power detector  100 G and further has a reception signal use first signal input terminal TIN 201 , local signal use second signal input terminal TIN 202 , branch circuits  201  and  202 , phase shifters  203  and  204 , a switching circuit  205 , and an N-port signal-IQ signal conversion circuit  206 . 
     Here, “3-port” means the three ports obtained by adding one port of the output terminal to the power detector  100 G of the branch circuit  201  to the two ports of the reception signal use first signal input terminal TIN 201  and the local signal use second signal input terminal TIN 202 . 
     Note that, in the demodulator of FIG. 16, the generating means is comprised by the branch circuits  201  and  202 , phase shifters  203  and  204 , and the switching circuit  205 . 
     In this 3-port demodulator  200 , a reception signal RS input to the input terminal TIN 201  is input to the branch circuit  201  and branched to two signals. One branched signal is input to the power detector  100 G. 
     Also, a local signal LS input to the input terminal TIN 202  is input to the branch circuit  202  and branched to two signals. One branched signal is input to the phase shifter  203 , given a phase shift e and then input to the switching circuit  205 . The other signal branched at the branch circuit  202  is input to the phase shifter  204 , given the phase shift θ, and then input to the switching circuit  205 . Then, the signal subjected to a phase shift function by the phase shifter  203  and the phase shifter  204  is sequentially changed over by the switching circuit  205  and supplied to the branch circuit  201 . 
     The signal input to the branch circuit  201  is branched to two signals to be supplied to the power detector  100 G and the input terminal TIN 201 . 
     In the power detector  100 , the amplitude component of the input signal is detected and supplied to the conversion circuit  206 . Then, in the conversion circuit  206 , the input signal is converted to an in-phase signal (I) and a quadrature signal (Q) as demodulated signals and output. 
     According to the 3-port demodulator  200 , the power detector  100 G can be easily used for broadband applications, so it can be applied to a system for which a multi-band or wide band characteristic is required. Further, demands for raising the frequency can be coped with. 
     Also, the power detector  100 G operates in a linear region, so demodulation is possible also by a low local signal power, and a low distortion demodulation is possible. 
     FIG. 17 is a circuit diagram of an example of the configuration of a time division 4-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     This 4-port demodulator  300  is comprised by using two high frequency power detectors  100 G- 1  and  100 G- 2  and further has a reception signal use input terminal TIN 301 , local signal use input terminal TIN 302 , switching circuits  301  and  302 , branch circuits  303  and  304 , phase shifter  305 , and N-port signal-IQ signal conversion circuit  306 . 
     Here, “4-port” means the four ports obtained by adding the two ports of the output terminal to the power detector  100 G- 1  of the branch circuit  303  and the output terminal to the power detector  100 G- 2  of the branch circuit  304  to the two ports of the reception signal use first input terminal TIN 301  and the local signal use second input terminal TIN 302 . 
     Note that, in the demodulator of FIG. 17, the generating means is comprised by the switching circuits  301  and  302 , branch circuits  303  and  304 , and the phase shifter  305 . 
     In this 4-port demodulator  300 , the reception signal RS input to the input terminal TIN 301  is input to the branch circuit  303  via the high speed switching circuit  301  and branched to two signals. One branched signal is input to the power detector  100 G- 1 , and the other signal is input to the phase shifter  305 . 
     In the phase shifter  305 , a phase shift e is given to the reception signal of the branch circuit  303 , and a signal subjected to the phase shift action is input to the branch circuit  304  and branched to two signals. At the branch circuit  304 , one branched signal is input to the power detector  100 G- 2 , while the other signal is supplied to the high speed switching circuit  302 . 
     Also, the local signal LS input to the input terminal TIN 302  is input via the high speed switching circuit  302  to the branch circuit  304  and branched to two signals. One branched signal is input to the power detector  100 G- 2 , while the other signal is input to the phase shifter  305 . 
     At the phase shifter  305 , a phase shift θ is given to the local signal by the branch circuit  304 , while the signal subjected to the phase shift action is input to the branch circuit  303  and branched to two signals. At the branch circuit  303 , one branched signal is input to the power detector  100 G- 1 , while the other signal is supplied to the high speed switching circuit  301 . 
     The power detector  100 G- 1  is supplied with the reception signal and the local signal given the phase shift θ. At the power detector  100 G- 1 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  306 . 
     Also, the power detector  100 G- 2  is supplied with the local signal and the reception signal given the phase shift θ. At the power detector  100 G- 2 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  306 . 
     Then, at the transform circuit  306 , the input signal is converted to the in-phase signal (I) and quadrature signal (Q) as the demodulated signals and output. 
     According to the present 4-port demodulator, similar effects to those of the 3-port demodulator can be obtained. 
     FIG. 18 is a circuit diagram of an example of the configuration of a 5-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     This 5-port demodulator  400  is comprised by using three high frequency power detectors  100 G- 1 ,  100 G- 2 , and  100 G- 3  and further has a reception signal use first signal input terminal TIN 401 , local signal use second signal input terminal TIN 402 , coupler  401 , branch circuits  402  and  403 , phase shifter  404 , and N-port signal-IQ signal conversion circuit  405 . Here, “5-port” means the five ports obtained by adding three ports of the output terminal to the power detector  100 G- 1  of the coupler  401 , the output terminal to the power detector  100 G- 2  of the branch circuit  402 , and the output terminal to the power detector  100 G- 3  of the branch circuit  403  to the two ports of the reception signal use input terminal TIN 401  and local signal use input terminal TIN 402 . 
     Note that, in the demodulator of FIG. 18, the generating means is comprised of the coupler  401 , branch circuits  402  and  403 , and the phase shifter  404 . 
     In this 5-port demodulator  400 , the reception signal RS input to the input terminal TIN 401  is input to the branch circuit  402  by the coupler  401 , and one part thereof is input to the power detector  100 G- 1 . 
     The reception signal input to the branch circuit  402  is branched to two signals. One branched signal is input to the power detector  100 G- 2 , while the other signal is input to the phase shifter  404 . 
     In the phase shifter  404 , a phase shift θ is given to the reception signal by the branch circuit  402 , and the signal subjected to the phase shift action is input to the branch circuit  403  and branched to two signals. At the branch circuit  403 , one branched signal is input to the power detector  100 G- 3 , while the other signal is supplied to the input terminal TIN 402 . 
     Also, the local signal LS input to the input terminal TIN 402  is input to the branch circuit  403  and branched to two signals. One branched signal is input to the power detector  100 G- 3 , while the other signal is input to the phase shifter  404 . 
     At the phase shifter  404 , a phase shift e is given to the local signal by the branch circuit  403 , and the signal subjected to the phase shift action is input to the branch circuit  402  and branched to two signals. At the branch circuit  402 , one branched signal is input to the power detector  100 G- 2 , while the other signal is supplied to the coupler  401 . 
     The power detector  100 G- 1  is supplied with the reception signal. At the power detector  100 G- 1 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  405 . 
     The power detector  100 G- 2  is supplied with the reception signal and the local signal given the phase shift θ. At the power detector  100 G- 2 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  405 . 
     Also, the power detector  100 G- 3  is supplied with the local signal and the reception signal given the phase shift θ. At the power detector  100 G- 3 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  405 . 
     Then, at the conversion circuit  405 , the input signal is converted to the in-phase signal (I) and quadrature signal (Q) as the demodulated signals and output. 
     According to the present 5-port demodulator, similar effects to those by the 3-port demodulator can be obtained. 
     FIG. 19 is a circuit diagram of an example of the configuration of a 6-port demodulator to which the high frequency power detector according to the present invention can be applied. 
     This 6-port demodulator  500  is comprised by using four high frequency power detectors  100 G- 1 ,  100 G- 2 ,  100 G- 3 , and  100 G- 4  and further has a reception signal use first signal input terminal TIN 501 , local signal use second signal input terminal TIN 502 , couplers  501  and  502 , branch circuits  503  and  504 , a phase shifter  505 , and an N-port signal-IQ signal conversion circuit  506 . 
     Here, “6-port” means the six ports obtained by adding the four ports of the output terminal to the power detector  100 G- 1  of the coupler  501 , the output terminal to the power detector  100 G- 2  of the branch circuit  502 , the output terminal to the power detector  100 G- 3  of the branch circuit  503 , and the output terminal to the power detector  100 G- 4  of the coupler  502  to the two ports of the reception signal use input terminal TIN 501  and the local signal use input terminal TIN 502 . 
     Note that, at the demodulator of FIG. 19, the generating means is comprised of the couplers  501  and  502 , branch circuits  503  and  504 , and the phase shifter  505 . 
     In this 6-port demodulator  500 , the reception signal RS input to the input terminal TIN 501  is input to the branch circuit  503  by the coupler  501 , and one part thereof is input to the power detector  100 G- 1 . 
     The reception signal input to the branch circuit  503  is branched to two signals. One branched signal is input to the power detector  100 G- 2 , and the other signal is input to the phase shifter  505 . 
     At the phase shifter  505 , a phase shift θ is given to the reception signal by the branch circuit  503 , and a signal subjected to the phase shift function is input to the branch circuit  504  and branched to two signals. At the branch circuit  504 , one branched signal is input to the power detector  100 G- 3 , and the other signal is input to the coupler  502 . 
     At the coupler  502 , the signal is supplied to the signal input terminal TIN 502  by the coupler  501 . 
     Also, the local signal LS input to the signal input terminal TIN 502  is input to the branch circuit  504  by the coupler  502 , and one part thereof is input to the power detector  100 G- 4 . 
     The local signal input to the branch circuit  504  is branched to two signals. One branched signal is input to the power detector  100 G- 3 , while the other signal is input to the phase shifter  505 . 
     At the phase shifter  505 , a phase shift θ is given to the local signal by the branch circuit  504 , and the signal subjected to the phase shift action is input to the branch circuit  503  and branched to two signals. At the branch circuit  503 , one branched signal is input to the power detector  100 G- 2 , and the other signal is supplied to the coupler  501 . 
     The power detector  100 G- 1  is supplied with only the reception signal. At the power detector  100 G- 1 , the amplitude component of the supplied reception signal is detected and supplied to the conversion circuit  506 . 
     The power detector  100 G- 2  is supplied with the reception signal and the local signal given the phase shift θ. At the power detector  100 G- 2 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  506 . 
     Also, the power detector  100 G- 3  is supplied with the local signal and the reception signal given the phase shift θ. At the power detector  100 G- 3 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  506 . 
     Also, the power detector  100 G- 4  is supplied with only the local signal. At the power detector  100 G- 4 , the amplitude component of the supplied signal is detected and supplied to the conversion circuit  506 . 
     Then, at the conversion circuit  505 , the input signal is converted to the in-phase signal (I) and quadrature signal (Q) as the demodulated signals and output. 
     According to the present 6-port demodulator, effects similar to those by the 3-port demodulator can be obtained. 
     INDUSTRIAL APPLICABILITY 
     As described above, according to the power detector of the present invention and the demodulator using the same, since they can be comprised on a semiconductor process suited for a high frequency such as GaAs in comparison with a detector using a silicon Schottky diode, they are suited for monolithic structures. Accordingly, a detector of a small size, low cost, and suitability for broadband high frequency operation can be realized. Also, the power detector of the present invention can realize a high performance power detector excellent in linearity of the detection characteristic, having a small fluctuation of detection characteristic relative to the bias fluctuation, having a small fluctuation of detection characteristic relative the FET threshold voltage fluctuation, and having a small DC offset. Also, the power detector of the present invention has a balance output, so when the latter stage circuit has a balance input, connection becomes easy.