Patent Publication Number: US-11652294-B2

Title: Dual-band antenna

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 16/134,519, filed on Sep. 18, 2018, which is a continuation of International Application No. PCT/CN2017/072085, filed on Jan. 22, 2017. All of the aforementioned patent applications are hereby incorporated by reference in their entireties. 
    
    
     TECHNICAL FIELD 
     The present application relates to the field of wireless communications, and in particular, to a coaxial dual-band antenna that can be used in a dual-band parabolic antenna. 
     BACKGROUND 
     With rapid development of wireless communications technologies, a transmission capacity in microwave point-to-point communication continuously increases, and an E-band (71 to 76 GHz, 81 to 86 GHz) frequency band microwave device plays an increasingly important role in a base station backhaul network. However, because “rain fade” on an E-band frequency band electromagnetic wave is extremely severe, an E-band microwave single-hop distance is usually less than 3 kilometers. To increase the E-band microwave single-hop distance and reduce site deployment costs, a solution is provided, in which the E-band frequency band microwave device and another low frequency microwave device are cooperatively used. When there is relatively heavy rain, even if the E-band microwave device cannot normally work, the low frequency microwave device can still normally work. 
     A dual-band parabolic antenna is used in this solution, and a structure of the dual-band parabolic antenna is shown in  FIG.  1   . The dual-band parabolic antenna includes a primary reflector, a secondary reflector, a low frequency feed, and a high frequency feed. The high frequency feed is inserted into the low frequency feed, and the two feeds use a same axis, and form a coaxial dual-band antenna. The two feeds of the coaxial dual-band antenna share the primary reflector and the secondary reflector, and phase centers of the two feeds are overlapped at a focus of the secondary reflector, so as to implement a dual-band multiplexing function. 
     In the prior art, a low frequency feed of a coaxial dual-band antenna is usually in a shape of a large-mouth horn, and a dielectric pin needs to be inserted into a high frequency feed. Both the high frequency feed and the low frequency feed have problems that radiation efficiency is relatively low, and a gain cannot reach a gain level of a single-band antenna. 
     SUMMARY 
     Embodiments of the present application provide a coaxial dual-band antenna. A circular waveguide with an unchanged diameter or a circular waveguide with a small flare angle is used to replace a large-mouth horn-shaped waveguide and serves as a low frequency feed, so as to resolve problems that radiation efficiency of a high frequency feed and a low frequency feed in an existing coaxial dual-band antenna is relatively low, and a gain cannot reach a gain level of a single-band antenna. 
     According to a first aspect, a coaxial dual-band antenna is provided and includes a waveguide tube, a ring groove, a high frequency feed, and a dielectric ring, where the waveguide tube has a tubular structure, and is configured to transmit a first electromagnetic wave, the ring groove whose opening direction is the same as an output direction of the first electromagnetic wave is on a wall of the waveguide tube, and a frequency of the first electromagnetic wave is lower than a frequency of an electromagnetic wave transmitted by the high frequency feed; the high frequency feed is located in the waveguide tube, and has a same axis with the waveguide tube, and the first electromagnetic wave excites a transverse electric mode TE 11  in the waveguide tube; and; and the dielectric ring is filled between the waveguide tube and the high frequency feed, the dielectric ring has a multi-layer structure, and has a same axis with the waveguide tube, area sizes of planes that are at layers of the dielectric ring and that are perpendicular to the axis alternately change, and a height of the dielectric ring is less than a height of the waveguide tube. 
     The coaxial dual-band antenna provided in the embodiments of the present application excites the TE 11  mode of the first electromagnetic wave at a low frequency, and no high order mode is generated inside the waveguide tube. This avoids a transmission loss of a high order mode in the waveguide, and improves low frequency radiation efficiency of the dual-band antenna. In addition, because no high order mode is generated inside the waveguide tube, there is no need to worry about that the high frequency feed located in the waveguide tube affects electromagnetic field distribution of the high order mode. Therefore, a dielectric pin can be omitted, and high frequency radiation efficiency of the dual-band antenna can be improved. 
     With reference to the first aspect, in a first possible implementation of the first aspect, a height of the high frequency feed is the same as the height of the waveguide tube. 
     With reference to the first aspect, in a second possible implementation of the first aspect, a sum of a radius of an inner wall of the waveguide tube and a radius of an outer wall of the high frequency feed is greater than 1/π of a wavelength of the first electromagnetic wave, and a difference between the two radiuses is less than ½ of the wavelength of the first electromagnetic wave. In the embodiments, it can be ensured that only the TE 11  mode is exited in the antenna, and a mode of a higher order does not exist, and therefore, a transmission loss of a high order mode in a waveguide is avoided. 
     With reference to the first aspect, or the first or the second possible implementation of the first aspect, in a third possible implementation of the first aspect, a difference between a radius of the ring groove and the radius of the inner wall of the waveguide tube is ⅛ of the wavelength of the first electromagnetic wave. 
     With reference to the third possible implementation of the first aspect, in a fourth possible implementation of the first aspect, a depth of the ring groove is between ⅕ and ¼ of the wavelength of the first electromagnetic wave, and a width of the ring groove is ⅛ of the wavelength of the first electromagnetic wave. 
     Size requirements of the ring groove are provided in the foregoing two implementations. A high order mode excited by a ring groove meeting the size requirements may be overlaid with the TE 11  mode, so that a beam width of the first electromagnetic wave on an E plane is consistent with that on an H plane, and radiation efficiency of the first electromagnetic wave is maximized. 
     With reference to any one of the first aspect, or the first to the fourth possible implementations of the first aspect, in a fifth possible implementation of the first aspect, an outer wall at only one of two adjacent layers of the dielectric ring is connected to the inner wall of the waveguide tube, and an inner wall at the layer of the dielectric ring is connected to the outer wall of the high frequency feed. This can implement sealing and waterproof functions, and can fasten the high frequency feed. 
     With reference to any one of the first aspect, or the first to the fifth possible implementations of the first aspect, in a sixth possible implementation of the first aspect, a layer that is of the dielectric ring and that is farthest from an output plane of the waveguide tube is not connected to the waveguide tube and the high frequency feed at a same time. This can reduce reflection of the first electromagnetic wave on the dielectric ring, and improve radiation efficiency. 
     With reference to the sixth possible implementation of the first aspect, in a seventh possible implementation of the first aspect, a height of each layer of the dielectric ring is ¼ of the wavelength of the first electromagnetic wave. 
     With reference to the sixth or the seventh possible implementation of the first aspect, in an eighth possible implementation of the first aspect, a relative dielectric constant of the dielectric ring is between 2 and 4. 
     The height of each layer of the dielectric ring and the relative dielectric constant are described in the foregoing two implementations. The height of each layer of the dielectric ring and the relative dielectric constant enable characteristic impedance of the coaxial dual-band antenna and wave impedance of free space to match each other, and improve the radiation efficiency. 
     The coaxial dual-band antenna provided in the present application excites a TE 11  mode of a first electromagnetic wave at a low frequency, and no high order mode is generated inside a waveguide tube. This avoids a transmission loss of a high order mode in the waveguide, and improves low frequency radiation efficiency of the dual-band antenna. In addition, because no high order mode is generated inside the waveguide tube, there is no need to worry about that a high frequency feed located in the waveguide tube affects electromagnetic field distribution of the high order mode. Therefore, a dielectric pin can be omitted, and high frequency radiation efficiency of the dual-band antenna can be improved. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG.  1    is a schematic structural diagram of an existing dual-band parabolic antenna; 
         FIG.  2    is a schematic structural diagram of an existing coaxial dual-band antenna: 
         FIG.  3  ( a )  is a schematic structural diagram of a coaxial dual-band antenna according to an embodiment of the present application; 
         FIG.  3  ( b )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  3  ( c )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  4  ( a )  is a distribution diagram of an electric field of a TE 11  mode in a coaxial dual-band antenna according to an embodiment of the present application; 
         FIG.  4  ( b )  is a distribution diagram of an electric field of a TM 11  mode in a coaxial dual-band antenna according to an embodiment of the present application; 
         FIG.  4  ( c )  is a distribution diagram of an electric field obtained after a TE 11  mode and a TM 11  mode in a coaxial dual-band antenna are overlaid according to an embodiment of the present application; 
         FIG.  5  ( a )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  5  ( b )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  6  ( a )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  6  ( b )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  7  ( a )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  7  ( b )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; 
         FIG.  8  ( a )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application; and 
         FIG.  8  ( b )  is a schematic structural diagram of a coaxial dual-band antenna according to another embodiment of the present application. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     The following describes the technical solutions in the embodiments of this application with reference to the accompanying drawings in the embodiments of this application. 
     A structure of an existing coaxial dual-band antenna is show in  FIG.  2   . A low frequency feed  201  of the coaxial dual-band antenna is a large-mouth horn-shaped waveguide, a high frequency feed  202  is included in the waveguide, and a dielectric pin  203  is inserted into the high frequency feed  202 . The horn-shaped waveguide is used to facilitate matching between characteristic impedance of the waveguide and wave impedance of free space, so as to reduce reflection. As a radius of the waveguide increases, a high order mode is excited, and the high order mode and a transverse electric mode TE 11  take effect, so that a beam width of an output electromagnetic wave on an E plane is consistent with that on an H plane, and a best gain effect is achieved. The E plane is a plane including a direction in which an electric field is located and a direction with highest radiation intensity, and the H plane is a plane including a direction in which a magnetic field is located and the direction with the highest radiation intensity. However, the high order mode is generated inside the large-mouth horn-shaped waveguide, and a transmission loss in the waveguide is relatively large. Therefore, low frequency radiation efficiency of the dual-band antenna is relatively low. 
     The high frequency feed is metallic, and affects electromagnetic field distribution of the high order mode. Therefore, the high frequency feed cannot directly extend to an aperture of the large-mouth horn-shaped waveguide, and a dielectric pin needs to guide a phase center of the high frequency feed to the aperture of the large-mouth horn-shaped waveguide. However, processing of the dielectric pin is uneasy, and a loss of the dielectric pin is relatively large. Therefore, a high frequency gain of the dual-band antenna cannot reach a level of a single-band antenna, either. 
     An embodiment of the present application provides a coaxial dual-band antenna. As shown in  FIG.  3  ( a ) , the antenna includes a waveguide tube  301 , a ring groove  302 , a high frequency feed  303 , and a dielectric ring  304 . 
     The waveguide tube  301  has a tubular structure, and is configured to transmit a first electromagnetic wave, the ring groove  302  whose opening direction is the same as an output direction of the first electromagnetic wave is on a wall of the waveguide tube  301 , and a frequency of the first electromagnetic wave is lower than a frequency of an electromagnetic wave transmitted by the high frequency feed  303 . 
     The high frequency feed  303  is located in the waveguide tube  301 , and has a same axis with the waveguide tube  301 , and the first electromagnetic wave excites a transverse electric mode TE 11  in the waveguide tube  301 . 
     The dielectric ring  304  is filled between the waveguide tube  301  and the high frequency feed  303 . The dielectric ring  304  has a multi-layer structure, and has a same axis with the waveguide tube  301 . Area sizes of planes that are at layers of the dielectric ring  304  and that are perpendicular to the axis alternately change. A height of the dielectric ring  304  is less than a height of the waveguide tube  301 . 
     Optionally, a height of the high frequency feed  303  is the same as the height of the waveguide tube  301 . It should be understood that, it is also feasible if the height of the high frequency feed is slightly less than the height of the waveguide tube. 
     In this embodiment of the present application, a waveguide tube excites a TE 11  mode of a first electromagnetic wave at a low frequency, and no high order mode is generated inside the waveguide tube. This avoids a transmission loss of a high order mode in the waveguide, and improves low frequency radiation efficiency of a dual-band antenna. In addition, because no high order mode is generated inside the waveguide tube, there is no need to worry about that a high frequency feed located in the waveguide tube affects electromagnetic field distribution of the high order mode. Therefore, a dielectric pin can be omitted, and high frequency radiation efficiency of the dual-band antenna can be improved. 
     It should be understood that, in the coaxial dual-band antenna shown in  FIG.  3  ( a ) , an inner wall of the dielectric ring  304  is connected to an outer wall of the high frequency feed  303 . This is only a possible structure of the coaxial dual-band antenna provided in the present application. Provided that area sizes of planes that are at layers of the dielectric ring  304  and that are perpendicular to the axis alternately change, alternatively, as shown in  FIG.  3  ( b ) , in the antenna, an outer wall of the dielectric ring  304  may be connected to an inner wall of the waveguide tube  301 ; or, as shown in  FIG.  3  ( c ) , an inner wall at one or more layers of the dielectric ring  304  may be connected to an outer wall of the high frequency feed  303 , and an outer wall at a remaining layer of the dielectric ring is connected to an inner wall of the waveguide tube  301 . 
     It should be noted that, electromagnetic field distribution on a cross section of a waveguide is referred to as a propagation mode of the waveguide. Different propagation modes have different cut-off wavelengths, a mode without a cut-off wavelength or with a longest cut-off wavelength is referred to as a dominant mode or a base mode, and another mode with a shorter cut-off wavelength is referred to as a high order mode. A higher order of a propagation mode indicates a shorter cut-off wavelength. In this embodiment of the present application, the TE 11  mode is used as the base mode, and another mode with a cut-off wavelength shorter than that of the TE 11  mode is referred to as a high order mode. 
     It should be understood that, the waveguide tube provided in this embodiment of the present application may be in a shape of a cylinder, a rectangular tube, or the like. A mouth for outputting the first electromagnetic wave may be slightly expanded, provided that only the base mode of the first electromagnetic wave is excited in the coaxial dual-band antenna including the waveguide tube, the high frequency feed, the ring groove, and the dielectric ring. A wall of the waveguide tube is usually metallic. 
     Optionally, a sum of a radius of the inner wall of the waveguide tube  301  and a radius of the outer wall of the high frequency feed  303  is greater than 1/π of a wavelength of the first electromagnetic wave, a difference between the two radiuses is less than ½ of the wavelength of the first electromagnetic wave, and a frequency of the first electromagnetic wave is lower than a frequency of an electromagnetic wave transmitted by the high frequency feed  303 . 
     Specifically, the coaxial waveguide including the high frequency feed  303  and the waveguide tube  301  in the present application is used as an example. Cut-off wavelengths of the first electromagnetic wave in different modes are related to an outer radius a of an inner waveguide (the radius of the outer wall of the high frequency feed  303 ) and an inner radius b of an outer waveguide (the radius of the inner wall of the waveguide tube  301 ) in the coaxial waveguide. A correspondence is listed in Table 1. 
     
       
         
           
               
               
             
               
                 TABLE 1 
               
               
                   
               
             
            
               
                 Propagation mode 
                 Cut-off wavelength 
               
               
                 TEM 
                 No cut-off wavelength 
               
               
                 TE 11   
                 π × (b + a) 
               
               
                 TM m1  (m = 0, 1, 2 . . . ) TE 01   
                 2 × (b − a) 
               
               
                 TE 21   
                 π × (b + a)/2 
               
               
                 TE m1  (m = 3, 4, 5 . . . ) 
                 π × (b + a)/m 
               
               
                 TM m2 , TE 02   
                 b − a 
               
               
                 TM mn  (n = 3, 4, 5 . . . ), TE 0n  (n = 3, 4, 5 . . . ) 
                 2 × (b − a)/n 
               
               
                   
               
            
           
         
       
     
     If the wavelength of the first electromagnetic wave is λ, it can be learned from Table 1 that the first electromagnetic wave may excite the TE 11  mode if the coaxial waveguide meets a condition in which (b+a)&gt;λ/π and (b−a)&lt;λ/2. If b in the coaxial waveguide becomes larger, and consequently (b−a)&gt;λ/2 and (b+a)&lt;2λ/π, the first electromagnetic wave may excite modes such as TE 11 , TM m1 , and TE 01  in theory. However, a continuous tangential component needs to be ensured when an electromagnetic field mode changes, that is, m needs to be consistent. Therefore, only two modes actually exist: TE 11  and TM 11 . As the inner radius b of the outer waveguide in the coaxial waveguide increases, more modes exist gradually. 
     It should be noted that, a transverse electromagnetic mode (TEM) may also exist in the coaxial waveguide, and no cut-off wavelength exists in this mode or a cut-off wavelength in this mode is infinitely long. However, before being excited in the coaxial dual-band antenna, the TEM mode is suppressed in a symmetrical feeding manner. Therefore, this mode is not considered in this embodiment of the present application. 
     Further, as shown in  FIG.  4  ( a ) , only the TE 11  mode exists in the waveguide tube, and electric field distribution of the TE 11  mode in the waveguide tube is non-uniform, that is, electric field distribution of the first electromagnetic wave is non-uniform. Therefore, a beam width of the first electromagnetic wave on the E plane is inconsistent with that on the H plane. For the foregoing problem, in this embodiment of the present application, the ring groove  302  whose opening direction is the same as the output direction of the first electromagnetic wave is excavated on the wall of the waveguide tube  301 , a high order mode is excited by using discontinuity of the wall of the waveguide tube  301 , and the high order mode is used to make the electric field distribution of the TE 11  mode become uniform. A depth and a width of the ring groove  302  and a distance from the ring groove  302  to the inner wall of the waveguide tube  301  all affect an order and amplitude of the high order mode. 
     Optionally, a difference between a radius of the ring groove  302  and the radius of the inner wall of the waveguide tube  301  is ⅛ of the wavelength of the first electromagnetic wave. The depth of the ring groove  302  is between ⅕ and ¼ of the wavelength of the first electromagnetic wave, and the width of the ring groove  302  is ⅛ of the wavelength of the first electromagnetic wave. Specifically, at a location that is on a wall plane at an output end of the waveguide tube and whose distance with the inner wall of the waveguide tube is ⅛ of the wavelength of the first electromagnetic wave, a ring whose width and depth meet the foregoing requirements is excavated on the wall, to form the ring groove  302 . The ring groove  302  causes discontinuity on a surface of the wall, so that the high order mode is excited. The location, the width, and the depth of the ring groove  302  meet the foregoing requirements, so that the high order mode TM 11  with an appropriate amplitude can be generated. Electric field distribution of the TM 11  mode is shown in  FIG.  4  ( b ) . As shown in  FIG.  4  ( c ) , the TE 11  mode and the TM 11  mode are overlaid, so that the electric field distribution of the first electromagnetic wave becomes uniform. Consequently, the beam width of the first electromagnetic wave on the F plane is consistent with that on the H plane, and a gain effect is maximized. 
     In addition, a large-mouth horn-shaped waveguide is omitted in this embodiment of the present application. Therefore, characteristic impedance of the coaxial dual-band antenna and wave impedance of free space cannot match each other by gradually changing the characteristic impedance at the output end of the waveguide tube by using a gradually increasing diameter of the waveguide tube. In this embodiment of the present application, impedance matching may be implemented in the following two manners. 
     (1) The dielectric ring  304  filled between the waveguide tube  301  and the high frequency feed  303  is used to implement impedance matching. The dielectric ring  304  has the multi-layer structure, and has the same axis with the waveguide tube  301 . The area sizes of the planes that are at the layers of the dielectric ring  304  and that are perpendicular to the axis alternately change. The height of the dielectric ring  304  is less than the height of the waveguide tube  301 . The structure of the dielectric ring  304  may be any structure shown in  FIG.  3  ( a ) ,  FIG.  3    ( b ), and  FIG.  3  ( c ) . 
     According to an impedance matching principle, when load impedance and characteristic impedance of a waveguide are inconsistent, to ensure that energy is transferred to a load and is not reflected back, a matching section is required between the load and the waveguide. When characteristic impedance Z 0  of the matching section meets the following formula, the characteristic impedance of the waveguide is equal to the load impedance after being converted by the matching section.
 
 Z   0 =√{square root over ( R   0   R   L )}(1)
 
     R 0  is the characteristic impedance of the waveguide, and R L  is the load impedance. 
     In this embodiment of the present invention, the load impedance is the wave impedance of the free space, and the characteristic impedance of the waveguide is the characteristic impedance of the coaxial dual-band antenna. The characteristic impedance of the waveguide tube can be changed by filling a dielectric in the waveguide tube. That is, the filled dielectric ring forms the matching section. However, if the waveguide tube is fully filled with the dielectric, in the waveguide tube, a sudden change of the characteristic impedance occurs on a contact surface between the dielectric and the air, and there is strong reflection. 
     The dielectric ring  304  used in the present application does not fully fill a gap between the waveguide tube  301  and the high frequency feed  303 , but uses the multi-layer structure having the same axis with the waveguide tube  301 . The area sizes of the planes that are at the layers of the dielectric ring  304  and that are perpendicular to the axis alternately change, to form a mixture of the dielectric and the air. Therefore, an equivalent relative dielectric constant is no longer equal to a relative dielectric constant of a material, and can be controlled and changed. A purpose of such control and change is to enable the characteristic impedance of the matching section to reach a value obtained by means of calculation by using the foregoing formula. 
     Optionally, a height of each layer of the dielectric ring  304  is ¼ of the wavelength of the first electromagnetic wave, and the first electromagnetic wave is a low frequency electromagnetic wave transmitted by the coaxial dual-band antenna. 
     Optionally, in a structure shown in  FIG.  5  ( a )  or  FIG.  5  ( b ) , an outer wall at only one of two adjacent layers of the dielectric ring  304  is connected to the inner wall of the waveguide tube  301 , and an inner wall at the layer of the dielectric ring  304  is connected to the outer wall of the high frequency feed  303 . In this way, inner walls at multiple layers of the dielectric ring  304  are connected to the outer wall of the high frequency feed  303 , and outer walls at the multiple layers of the dielectric ring  304  are connected to the inner wall of the waveguide tube  301 . This can implement air sealing and waterproof functions, and can fasten the high frequency feed  303  in between. Consequently, the coaxial dual-band antenna not only can be applied to satellite communication, but also is applicable to the ground. Other than the layers of the dielectric ring that are connected to both the waveguide tube  301  and the high frequency feed  303 , spacing between an inner wall and an outer wall at another layer of the dielectric ring  304  needs to be designed and optimized according to the foregoing equivalent dielectric constant principle. 
     Optionally, a layer that is of the dielectric ring  304  and that is farthest from an output plane of the waveguide tube  301  is not connected to the waveguide tube  301  and the high frequency feed  303  at a same time, so as to reduce reflection of the first electromagnetic wave. The layer that is of the dielectric ring and that is farthest from the output plane is a bottom layer of the dielectric ring shown in  FIG.  5  ( a )  and  FIG.  5  ( b ) . 
     A dielectric material whose relative dielectric constant is between 2 and 4 may be used for the dielectric ring in this embodiment of the present application, for example, polycarbonate, polystyrene, and polytetrafluorethylene. A specific material is not limited in this embodiment of the present application. 
     After the material is determined, spacing between an inner wall and an outer wall at each layer of the dielectric ring  304  is further related to the wavelength of the first electromagnetic wave. The following provides a specific embodiment in which the frequency of the first electromagnetic wave is 18 GHz. It is assumed that polycarbonate whose relative dielectric constant is 2.8 is used to prepare the dielectric ring, the radius of the inner wall of the waveguide tube is R, and the dielectric ring has six layers. As shown in  FIG.  5  ( a ) , radius lengths of layers of the dielectric ring alternately change from top to bottom radiuses of outer walls at the first layer, the third layer, and the fifth layer of the dielectric ring are R, a radius of an outer wall at the second layer of the dielectric ring is 0.78 R, a radius of an outer wall at the fourth layer of the dielectric ring is 0.7 R, and a radius of an outer wall at the sixth layer of the dielectric ring is 0.7 R. The characteristic impedance of the matching section can meet formula (1) by using the dielectric ring with the foregoing sizes, so that the characteristic impedance of the coaxial dual-band antenna and the wave impedance of the free space match each other, electromagnetic wave reflection is reduced, and radiation efficiency is improved. 
     (2) Multiple metal rings  601  are disposed in the waveguide tube to implement impedance matching. The metal rings form a matching section. A possible structure is shown in  FIG.  6  ( a ) , and an inner wall of each metal ring  601  is connected to the outer wall of the high frequency feed  303 . Equivalent inductance and equivalent capacitance of each metal ring  601  may be changed by changing a radius of each metal ring  601  and spacing between the metal rings  601 , so that characteristic impedance of the matching section reaches a value obtained by means of calculation by using formula (1). 
     Optionally, a dielectric layer  602  may further be filled at a location that is inside the waveguide tube  301  and that is close to the output plane. As shown in  FIG.  6  ( b ) , an inner wall of the dielectric layer  602  is connected to the outer wall of the high frequency feed  303 , and an outer wall of the dielectric layer  602  is connected to the inner wall of the waveguide tube  301 . This can implement air sealing and waterproof functions, and can fasten the high frequency feed. A hard material may be used for the dielectric layer  602 , and a specific material is not limited in the present application. 
     It should be understood that,  FIG.  6  ( a )  and  FIG.  6  ( b )  show only possible structures in this embodiment of the present application. As shown in  FIG.  7 ( a )  and  FIG.  7  ( b ) , outer walls of the metal rings  601  may be connected to the inner wall of the waveguide tube  301 , to form a matching section. Alternatively, as shown in  FIG.  8  ( a )  and  FIG.  8  ( b ) , outer walls of some metal rings  601  are connected to the inner wall of the waveguide tube  301 , and inner walls of the other part of metal rings  601  are connected to the outer wall of the high frequency feed  303 , to form a matching section. A specific implementation is not limited in this embodiment of the present application. 
     The coaxial dual-band antenna provided in the present application has the following advantages: A waveguide tube  301  excites a TE 11  mode of a first electromagnetic wave at a low frequency, and no high order mode is generated inside the waveguide tube  301 . This avoids a transmission loss of a high order mode in the waveguide tube  301 , and improves low frequency radiation efficiency of the dual-band antenna. In addition, because no high order mode is generated inside the waveguide tube  301 , there is no need to worry about that a high frequency feed  303  located in the waveguide tube  301  affects electromagnetic field distribution of the high order mode. Therefore, a dielectric pin can be omitted, and high frequency radiation efficiency of the dual-band antenna can be improved. In addition, according to a design of a ring groove  302  and a dielectric ring  304 , a beam width of the first electromagnetic wave on an E plane can be consistent with that on an H plane, and characteristic impedance of the coaxial dual-band antenna and wave impedance of free space can match each other. 
     The foregoing descriptions are merely specific implementations of this application, but are not intended to limit the protection scope of this application. Any variation or replacement readily figured out by a person skilled in the art within the technical scope disclosed in this application shall fall within the protection scope of this application. Therefore, the protection scope of this application shall be subject to the protection scope of the claims.