Patent Publication Number: US-7710063-B2

Title: Electric power converter

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   This application claims priority from Japanese Patent Application Serial No. 2006-075094, filed Mar. 17, 2006, which is incorporated herein in its entirety by reference. 
   TECHNICAL FIELD 
   The invention relates in general to a type of electric power converter, and more particularly, a type of electric power converter for feeding driving electric power to a motor. 
   BACKGROUND 
   One known electric power converter is described in an “Electric power converter, and 2-power source system vehicle carrying same” in Japanese Kokai Patent Application No. 2006-25518. In this power converter, electric power is fed from plural power sources so as to drive a motor at high efficiency. 
   In the motor driving system described therein, plural power sources are connected in parallel to form a power source system. By feeding electric power to the motor from three or more potentials including a common potential, it is possible to drive the motor at any electric power allotment. In this way, it is possible to feed driving electric power to the motor from plural power sources simultaneously without having a DC-DC converter. As a result, it is possible to suppress loss and to realize high energy efficiency. 
   BRIEF SUMMARY OF THE INVENTION 
   One embodiment of an electric power converter for controlling feed voltages from plural power sources and driving a multi phase AC motor taught herein comprises an electric power conversion circuit and an electric power controller. The electric power controller is operable to control switching devices of the electric power conversion circuit to generate a driving voltage for driving at least a first phase of the motor by generating pulses from output voltages of the plural power sources and to generate a driving voltage for driving a different phase of the motor by generating pulses from an output voltage of only one of the plural power sources. 
   Electric power conversion systems are also taught herein. One example of a system for driving a multi phase AC motor including plural power sources where the plural power sources include at least a first power source and a second power source comprises an electric power converter configured to connect the first power source, the second power source and the motor. The electric power converter includes switches connected the plural power sources and operable to produce a first driving voltage for driving the motor by generating pulses from output voltages of the plural power sources and a switch connected to only one of the plural power sources and operable to produce a second driving voltage for driving the motor by generating pulses from an output voltage of the only one of the plural power sources. 
   Embodiments of a control method for an electric power converter for driving a multi phase AC motor using plural power sources are also taught herein. One method comprises generating voltage instruction values of each phase of the motor including a first voltage instruction value of a phase generating pulses from output voltages of the plural power sources, allotting the first voltage instruction value to respective voltage instruction values of each power source corresponding to an electric power allotment target, computing a modulation rate of an operation of a switch corresponding to each power source in the allotting step, correcting the modulation rate of the operation corresponding to each power source in the allotting step using a respective voltage of each power source, computing a modulation rate of an operation of a switch corresponding to a phase generating pulses from only one power source based on a voltage instruction value of the phase generating pulses from the only one power source, computing an offset voltage composed of a phase voltage of the phase generating pulses from only one power source and a phase voltage of the phase generating pulses from the output voltages of each of the plural power sources, amending a voltage instruction value of the phase generating pulses from the output voltages of the plural power sources, actuating a switch of the phase generating pulses from the output voltages of the plural power sources based on the modulation rate amended by the switch operation corresponding to the power sources and actuating a switch of the phase generating pulses from the only one power source based on the modulation rate of the phase generating pulses from the only one power source. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The description herein makes reference to the accompanying drawings wherein like reference numerals refer to like parts throughout the several views, and wherein: 
       FIG. 1  is circuit diagram illustrating components of an electric power converter according to the first embodiment of the invention; 
       FIG. 2  is a waveform illustrating the current waveform of a motor (U-phase current iu, V-phase current iv) when the motor is driven while the feed electric power is shared by the fuel cell and the battery in the first embodiment; 
       FIG. 3  is a block diagram illustrating components of the electric power conversion control system in the first embodiment; 
       FIG. 4  is a block diagram illustrating in detail the current control part shown in  FIG. 3 ; 
       FIG. 5  is a block diagram illustrating in detail the electric power control/modulation rate arithmetic operation part shown in  FIG. 3 ; 
       FIG. 6  is a block diagram illustrating the arithmetic operation in computing the modulation rate; 
       FIG. 7  is a diagram illustrating the input/output value in the modulation rate offset arithmetic operation unit of the electric power control/modulation rate arithmetic operation part shown in  FIG. 5 ; 
       FIG. 8  is a diagram illustrating the input/output values of the voltage offset compensation value arithmetic operation unit that computes the voltage offset compensation value; 
       FIG. 9  is a circuit diagram illustrating the U-phase in  FIG. 1 ; 
       FIG. 10  is a diagram illustrating the waveform of the sawtooth waves used in the PWM pulse generating part shown in  FIG. 3 ; 
       FIG. 11  is a waveform diagram illustrating generation of pulses of driving signal A and driving signal E by means of a sawtooth wave comparison; 
       FIG. 12  is a waveform diagram illustrating generation of pulses of driving signal D and driving signal C by means of a sawtooth wave comparison; 
       FIG. 13  is a waveform diagram illustrating an example of pulse generation attached with dead time Td; 
       FIG. 14  is a circuit diagram illustrating the components of the driving signal processor part shown in  FIG. 8 ; 
       FIG. 15  is a circuit diagram illustrating alternative components of the electric power converter shown in  FIG. 1 ; 
       FIG. 16  is a circuit diagram illustrating alternative components of the electric power converter shown in  FIG. 1 ; 
       FIG. 17  is a block diagram illustrating components of the electric power conversion control system according to a second embodiment of the invention; 
       FIG. 18  is a circuit diagram illustrating in additional detail the electric power converter shown in  FIG. 17 ; 
       FIG. 19  is a block diagram illustrating the modulation rate offset arithmetic operation part according to a third embodiment of the invention; 
       FIG. 20   a  is a phase current waveform illustrating the case when voltage offset compensation is not performed; 
       FIG. 20   b  is a phase current waveform illustrating the case wherein a voltage offset value is added; 
       FIG. 21  is a block diagram illustrating the modulation rate offset arithmetic operation part according to a fourth embodiment of the invention; 
       FIG. 22  is a block diagram illustrating the electric power control/modulation rate arithmetic operation part according to a fifth embodiment of the invention; 
       FIG. 23  is a block diagram illustrating the components of the electric power conversion control system according to a sixth embodiment of the invention; 
       FIG. 24  is a block diagram illustrating in detail the components of the torque control part shown in  FIG. 23 ; 
       FIG. 25  is a flow chart illustrating processing of the electric power controller in the sixth embodiment; 
       FIG. 26  is a diagram schematically illustrating the selection of the sign of the d-axis current and the selection of high frequency current; 
       FIG. 27  includes graphs illustrating an example of the result of electric power control in the sixth embodiment; 
       FIG. 28  is a block diagram illustrating the components of the electric power conversion control system in a seventh embodiment of the invention; 
       FIG. 29  is a block diagram illustrating in detail the torque control part shown in  FIG. 28 ; 
       FIG. 30  is a flow chart illustrating processing of the electric power controller in the seventh embodiment; 
       FIG. 31  is a diagram illustrating the braking device installed on the output shaft of the motor in the electric power conversion control system according to the eighth embodiment of the invention; 
       FIG. 32  is a block diagram illustrating in detail the torque control part in the eighth embodiment; 
       FIG. 33  is a block diagram illustrating the components of the electric power controller shown in  FIG. 32 ; 
       FIG. 34  is a diagram illustrating the clutch device installed on the output shaft of the motor of the electric power conversion control system according to a ninth embodiment of the invention; 
       FIG. 35  is a block diagram illustrating in detail the torque control part in the ninth embodiment; 
       FIG. 36  is a circuit diagram illustrating the electric power converter in a tenth embodiment of the invention; 
       FIG. 37  is a block diagram illustrating the components of the electric power conversion control system in the tenth embodiment; and 
       FIG. 38  is a circuit diagram illustrating an example of the power source in according to the first through tenth embodiments. 
   

   DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION 
   For the known electric power converter described above, switches are set for feeding electric power from plural power sources in each phase matched to the driving phase number of the motor. As a result, the number of switches is large, and the cost is increased. 
   In contrast, embodiments of the invention provide a type of electric power converter that, without using a DC-DC converter, uses and allots the electric power of plural power sources while reducing the overall volume and loss with fewer semiconductor elements. 
   Accordingly, one embodiment of the electric power converter for driving a multi-phase AC motor has a phase in which it is connected to plural power sources and generates and synthesizes pulses from the output voltages of plural power sources so as to drive the multi-phase AC motor and a phase in which it is connected to one DC power source and generates pulses from the output voltage of the power source so as to generate a driving voltage for the multi-phase AC motor. It allows use/allotment of the power of plural power sources with fewer semiconductor elements. 
   In the following, an explanation will be given regarding embodiments of the invention with reference to the figures. 
     FIG. 1  is a circuit diagram illustrating an electric power converter according to a first embodiment of the present invention. As shown in  FIG. 1 , electric power converter  12  has plural groups of switching devices for the various phases (U-phase, V-phase, W-phase) of motor  15 . 
   For both DC power source  11   a  and DC power source  11   b , the negative electrode side is connected to common negative electrode bus  16 . Common negative electrode bus  16  and the various phase terminals of motor  15  are connected via groups of semiconductor switches  17   a ,  18   a  and  19   a  and diodes  17   b ,  18   b  and  19   b , respectively, just like the lower arm of a conventional inverter. Regarding the connection between positive electrode bus  20  of DC power source  11   a  and the various phase terminals of motor  15 , for the V-phase and W-phase, connection is performed via groups of semiconductor switches  21   a ,  22   a  and diodes  21   b ,  22   b . For the U-phase, connection is performed via the groups of semiconductor switches  23   a ,  23   b  that can control bidirectional conduction. 
   Connection between positive electrode bus  24  of DC power source  11   b  and the U-phase terminal of motor  15  is realized via the group of two semiconductor switches  25   a ,  25   b . Connection between positive electrode bus  20  of DC power source  11   a  and common negative electrode bus  16  is performed via smoothing capacitor  26 , and connection between positive electrode bus  24  of DC power source  11   b  and common negative electrode bus  16  is performed by smoothing capacitor  27 . 
   This electric power converter  12  generates an output voltage applied on motor  15  as follows. For its U-phase, a voltage is generated based on the three potentials of common negative electrode bus  16 , positive electrode bus  20  of DC power source  11   a  and positive electrode bus  24  of DC power source  11   b . For the V-phase and W-phase, a voltage is generated based on the two potentials of common negative electrode bus  16  and positive electrode bus  20  of DC power source  11   a . The semiconductor switches set for phases of motor  15  are switching devices that generate the voltages output to the phases of motor  15 . From these potentials, one is selectively connected, and the necessary voltage is fed to motor  15  by changing the proportion of the connection time. 
   Electric power converter  12  has two functions. As one function, the three potential voltages of plural DC power sources are used, and the voltage needed for motor  15  is generated. As the other function, the electric power fed from DC power source  11   a  and DC power source  11   b  is set at any value. 
   For the former function, corresponding to the operation point of motor  15 , the AC voltage needed for motor  15  is generated from the DC voltage by a PWM. More specifically, from the 3-level voltage an AC voltage is generated by the PWM. 
   The latter function is described in more detail herein. For example, in the motor driving system using a fuel cell as DC power source  11   a  and a rechargeable battery as DC power source  11   b , from the viewpoint of efficiency and the response property of the fuel cell, etc., it is desirable that it be possible to set the proportion of electric power fed from the fuel cell and that from the battery at any value. Among other differences, embodiments of the invention differ from known converters in that it has heretofore been known to require a switch corresponding to the battery side of the V-phase and W-phase. 
   Here, the output voltage generated from the voltage of the fuel cell is (vuf, vvf, vwf), and the output voltage generated from the battery voltage is (vub, vvb, vwb)=(vub, 0, 0). Consequently, the electric power fed from the power source can be represented as follows:
 
 P =( vuf+vub,vvf+vvb,vwf+vwb )·( iu,iv,iw )=( vuf,vvf,vwf )·( iu,iv,iw )+( vub, 0,0)·( iu,iv,iw ).
 
On the right side, the first item is electric power Pf fed from DC power source  11   a:  
 
 Pf =( vuf,vvf,vwf )·( iu,iv,iw ); and
 
the second item is electric power Pb fed from DC power source  11   b:  
 
 Pb =( vub, 0,0)·( iu,iv,iw ).
 
   The switches connected to DC power source  11   b  are only U-phase switches  25   a ,  25   b . However, as can be seen from the formulas above, by adjusting (vuf, vvf, vwf) and (vub, 0, 0), it is possible to adjust Pf and Pb to any proportion. That is, in order to drive the motor, it is only required that the final values of (iu, iv, iw) be in agreement with the instruction values. By adjusting vuf, vvf, vwf and vub, 0, 0 in the range where iu, iv and iw agree with each other, it is possible to obtain any electric power Pf and electric power Pb. 
     FIG. 2  illustrates the motor current waveform (U-phase current iu, V-phase current iv) when the motor is driven while the feed electric power is shared by the fuel cell and the battery in the first embodiment. From  FIG. 2  it can be seen that the current can be controlled normally. 
   As explained above, in the present example, the number of switches is reduced over what has been known. Also, by means of switch devices  21   a ,  22   a ,  21   b ,  22   b  connected to DC power source  11   a  of the V-phase and W-phase, a backward voltage rating is not required. As a result, device elements  21   b ,  22   b  can be diodes. That is, as it is possible to reduce the number of elements and the number of reverse blocking function elements, it is possible to cut the cost and to reduce the size and weight. 
     FIG. 3  is a block diagram illustrating an electric power converter according to the first embodiment. As shown in  FIG. 3 , electric power conversion control system  10  has plural (two in this example) DC power sources  11   a ,  11   b  supplying respective power Pa, Pb, electric power converter (or conversion device)  12 , torque controller  13  and electric power controller  14 . The necessary voltage is fed from electric power converter  12  to a multi-phase motor  15 . Here, motor  15  is a 3-phase AC motor receiving power Pm. Torque controller  13  and electric power controller  14  generally comprise a microcomputer including a central processing unit (CPU), input and output ports (I/O), random access memory (RAM), keep alive memory (KAM), a common data bus and read-only memory (ROM) as an electronic storage medium for executable programs and certain stored values as discussed hereinafter. The various parts of the electric power controller  14  could be, for example, implemented in software as the executable programs, or could be implemented in whole or in part by separate hardware in the form of one or more integrated circuits (IC). 
   As shown in  FIG. 3 , instruction value id* for the direct-axis (d-axis) current of motor  15  and instruction value iq* for the quadrature-axis (q-axis) current are computed from torque instruction value Te* and motor rotation speed ω in torque controller  13 . Torque instruction value Te* is externally provided. According to one embodiment of torque controller  13 , a previously-prepared map is stored that takes Te* and ω as axes and provides id*, iq* as output values. 
   As also shown in  FIG. 3 , electric power controller  14  has current control part  28 , electric power controlling/modulating arithmetic operation part  29 , pulse width modulation (PWM), pulse generating part  30 , driving signal processor part  31  and 3-phase/dq conversion part  32 . 
   Current control part  28  controls the current from d-axis current-instruction value id*, q-axis current instruction value iq*, d-axis current value id and q-axis current value iq of motor  15  obtained from 3-phase/dq conversion part  32  so that the instruction values are in agreement with the measured values. By means of this control, voltage instruction values vu*, vv*, vw* of the various phases of the 3-phase AC motor  15  are output. 
     FIG. 4  is a block diagram illustrating the components of the current control part  28  shown in  FIG. 3 . Current control part  28  has control part  33  and dq/3-phase conversion part  34 . Control part  33  performs feedback control by means of proportional-integration (P-I) control so that d-axis current value id and q-axis current value iq follow d-axis current instruction value id* and q-axis current instruction value iq*, respectively. Control part  33  also outputs d-axis voltage instruction value vd* and q-axis voltage instruction value vq*. As explained above, d-axis current value id and q-axis current value iq are determined from U-phase current iu and V-phase current iv by means of 3-phase/dq conversion part  31 . 
   In current control part  28 , dq/3-phase conversion part  34  converts the d·q axis voltage instruction values to a 3-phase voltage instruction. More specifically, part  34  takes d-axis voltage instruction value vd* and q-axis voltage instruction value vq* as inputs, and it outputs U-phase voltage instruction value vu*, V-phase voltage instruction vv* and W-phase voltage instruction value vw* using angle θ. 
   Returning now to  FIG. 3 , electric power controlling/modulating arithmetic operation part  29  controls the electric power by using electric power distribution target values rto_pa, rto_pb fed from DC power source  11   a  and DC power source  11   b . The electric power distribution target values may be target values given from the outside. For example, they may be freely determined from the state of the vehicle, the electric power residual quantity of the DC power source, etc. The electric power distribution target values represent the proportions of the electric power of DC power source  11   a  and DC power source  11   b , and the electric power distribution target values rto_pa, rto_pb have the following relationship:
 
 rto   —   pa+rto   —   pb= 1.
 
   Because only the U-phase has electric power fed from plural power sources in this embodiment, however, in consideration of this feature adjustment is performed beforehand so that the target value of rto_pb in the U-phase is a multiple of the phase number with respect to the distribution target value of the power source. For example, assuming that the target of DC power source  11   a  is 0.7, when electric power can be distributed for all three phases, the target of DC power source  11   b  is 0.3. Now, in the present embodiment, in order to share the electric power that is only in the U-phase of DC power source  11   b , it is three times 0.3, that is, 0.9. On the other hand, for the U-phase of DC power source  11   a , the electric power share of DC power source  11   a  is set at 0.1 so that the rto_pa+rto_pb=1 is met. Such arithmetic operation may be performed beforehand when the distribution target is generated externally. 
   Consequently, when the electric power distribution target value on one side is obtained from the arithmetic relationship, it is possible to determine the electric power distribution target value on the other side. That is, regarding the input to electric power controlling/modulating arithmetic operation part  29 , for example, it may be only the electric power distribution target value rto_pa of DC power source  11   a  (see  FIG. 3 ), and electric power distribution target value rto_pb of DC power source  11   b  is computed based on the arithmetic relationship. 
   In the following, an explanation is given regarding the U-phase as the phase that generates pulses from plural power sources. 
     FIG. 5  is a block diagram illustrating in detail the electric power controlling/modulating arithmetic operation part  29  of  FIG. 3 . As shown in  FIG. 5 , electric power controlling/modulating arithmetic operation part  29  has subtractors  35 ,  36 , multiplier  37 , modulation rate arithmetic operation part  38  and modulation rate amendment part  39 . Subtractor  35  subtracts voltage offset compensation value v_ 0 * from input U-phase voltage instruction value vu* to determine voltage assignment value vu_ 0 *. Details of voltage offset compensation value v_ 0 * are explained later. Multiplier  37  multiplies electric power distribution target value rto_pa by voltage assignment value vu_ 0 * to determine voltage instruction value vu_a* for DC power source  11   a.    
   In the following, the instruction of the voltage generated from DC power source  11   a  will be called power source a portion voltage instruction, and the instruction of the voltage generated from DC power source  11   b  is called power-source b portion voltage instruction. 
   Voltage instruction value vu_a* on the side of DC power source  11   a  is determined by multiplying electric power distribution target value rto_pa by the result obtained by excluding voltage offset compensation value v_ 0 * from voltage instruction value vu* according to:
 
 vu   — 0 *=vu*−v   — 0*; and
 
 vu   —   a*=vu   — 0 *·rto   —   pa.  
 
   On the other hand, the voltage instruction value on the side of DC power source  11   b  is determined by subtracting voltage instruction value vu_a* on the side of DC power source  11   a  from voltage assignment value vu_ 0 * obtained from the control voltage of the motor current control by means of subtractor  36  according to:
 
 vu   —   b*=vu   — 0* −vu   —   a*.  
 
   The following explanation regarding computation of the modulation rate and generation of PWM pulses is for the U-phase. For the V-phase and W-phase, the modulation rate is computed only from voltage instruction values vv*, vw* and voltage Vdc_a of DC power source  11   a  connected to V- and W-phases. 
     FIG. 6  is a block diagram illustrating the arithmetic operation for computation the modulation rate. By means of multiplier  40 , 2/Vdc_a is multiplied by voltage instruction value vv* to determine modulation rate instruction value mv_a_c*.
   mv   —   a   —   c*=vv /( Vdc   —   a/ 2); and   mw   —   a   —   c*=vw */( Vdc   —   a/ 2). 
   Returning now to  FIG. 5 , modulation rate arithmetic operation means  38  generates instant modulation rate instruction values mu_a*, mu_b* as the normalized voltage instruction from voltage Vdc_a of DC power source  11   a  and voltage Vdc_b of DC power source  11   b . That is, modulation rate arithmetic operation means  38  has multipliers  41 ,  42 . Voltage instruction value vu_a* of the portion of power source a and voltage instruction value vu_b* of the portion of power source b of the U-phase are normalized by a value that is half their DC voltage, respectively, so that instant modulation rate instruction value mu_a* of the portion of power source a and instant modulation rate instruction value mu_b* of the portion of power source b are determined according to:
 
 mu   —   a*=vu   —   a */( Vdc   —   a/ 2); and
 
 mu   —   b*=vu   —   b */( Vdc   —   b/ 2).
 
     FIG. 7  is a diagram illustrating the input/output values in the modulation rate offset arithmetic operation unit  43  of the modulation rate correcting part  39  shown in  FIG. 5 . As shown in  FIG. 7 , modulation rate correcting part  39  uses modulation rate offset arithmetic operation unit  43  to allot the time width of the PWM period and to compute the final modulation rate instruction value so as to output the obtained modulation rate. 
   First, in modulation rate offset arithmetic operation unit  43 , from power source voltage Vdc_a of DC power source  11   a , power source voltage Vac_b of DC power source voltage source  11   b , and electric power distribution target value rto_pa of DC power source  11   a , the next modulation rate offsets ma_offset 0 , mb_offset 0  are computed according to: 
           ma_offset0   =            rto_pa   Vdc_a                   rto_pa   Vdc_a          +          rto_pb   Vdc_b                        and             mb_offset0   =            rto_pb   Vdc_b                   rto_pa   Vdc_a          +          rto_pb   Vdc_b                    
Here, electric power distribution target value rto_pb of DC power source  11   b  is computed using the formula rto_pb=1−rto_pa as discussed previously.
 
   Then, by means of adder  44  and adder  45 , modulation rate offsets ma_offset 0  and mb_offset 0  are added, respectively, to instant modulation rate instruction value-mu_a* of the portion of power source a and instant modulation rate instruction value mu_b* of the portion of power source b. The final modulation rate instruction values mu_a_c*, mu_b_c* are determined using the following formulas:
 
 mu   —   a   —   c*=mu   —   a*+ma _offset−1; and
 
 mu   —   b   —   c*=mu   —   b*+mb _offset−1.
 
     FIG. 8  is a diagram illustrating the input/output values in the voltage offset compensation arithmetic operation unit  46  that computes the voltage offset compensation value v_ 0 *. By means of voltage offset compensation arithmetic operation unit  46 , voltage offset compensation value v_ 0 * is computed using modulation rate offsets ma_offset 0  and mb_offset 0  and the power source voltage. 
   As seen from DC power source  11   a  alone, that is, when electric power distribution target value rto_pa=1, for both the V-phase and W-phase average value vu_out_ave 1  of output voltage vu_out for one period of the electrical angle is as follows:
 
 vu _out —   ave 1 =Vdc   —   a/ 2.
 
On the other hand, when electric power distribution target value rto_pa is set at a value different from 1, the average value becomes the sum of the voltage average value output from DC power source  11   a  and the voltage average value output from DC power source  11   b  according to:
 
 vu _out —   ave 2 =Vdc   —   a/ 2 ·ma _offset*+ Vdc   —   b/ 2 ·mb _offset*.
 
The difference in the output voltages Δvu_out_ave can be determined as follows:
 
Δ vu _out —   ave=Vdc   —   a/ 2 ·ma _offset*+ Vdc   —   b/ 2 ·mb _offset*− Vdc   —   a/ 2.
 
   Because the difference in the output voltages Δvu_out_ave becomes the offset voltage when the V-phase and W-phase are compared, the following voltage offset compensation value v_ 0 * is computed so that an offset current does not flow in motor  15  according to:
 
 v   — 0*=Δ vu _out —   ave.  
 
   In this embodiment, as explained above, because control is performed based on the value obtained by subtracting voltage offset compensation value v_ 0 * from U-phase voltage instruction value vu*, the phase voltage of the U-phase is in agreement with the phase voltage of the V-phase and W-phase, and a flow of offset current is prevented. 
     FIG. 9  is a circuit diagram illustrating the U-phase in  FIG. 1 , and  FIG. 10  illustrates the waveform of triangular waves used in the PWM pulse generating part  30  shown in  FIG. 3 . As shown in  FIG. 10 , carrier Ca for DC power source  11   a  is triangular wave carrier for generating PWM pulses that drive the various switching devices for output of voltage pulses from voltage Vdc_a of DC power source  11   a . Similarly, triangular wave carrier is set as carrier Cb for DC power source  11   b . For the two triangular wave carriers Ca, Cb, the upper limit is +1, the lower limit is −1, and there is a phase difference of 180°. Signals A-E that drive the various switching means of the U-phase are shown in  FIG. 9  as follows: 
   A: Signal for driving the switch  23   a  for conduction from DC power source  11   a  to the output terminal; 
   B: Signal for driving the switch  17   a  for conduction from the output terminal to the negative electrode; 
   C: Signal for driving the switch  23   b  for conduction from the output terminal to DC power source  11   a;    
   D: Signal for driving the switch  25   a  for conduction from DC power source  11   b  to the output terminal; and 
   E: Signal for driving the switch  25   b  for conduction from the output terminal to DC power source  11   b.    
   Next, an explanation is given for the pulse generation method when voltage pulses are output from DC power source  11   a . When PWM pulses are output from DC power source  11   a , driving signal A must be in the ON state. If there is a potential difference between the positive electrode of DC power source  11   a  and the positive electrode of DC power source  11   b , and power source voltage Vdc_a of DC power source  11   a  is higher than power source voltage Vdc_b of DC power source  11   b  (Vdc_a&gt;Vdc_b), when both driving signal A and driving signal E are ON, a short circuit current flows between the two positive electrodes. On the other hand, driving signal E is provided secure a return current course of DC power source  11   b . When PWM pulse is supplied by the DC power source  11   b , driving signal E is always an ON state. 
   For example, when driving signal A is switched from ON to OFF, and driving signal E is simultaneously switched from OFF to ON, since time is required for driving signal A to become fully OFF, an overlap occurs with the ON state of driving signal E. Both are ON for a time, a short circuit current flows, and the heat generated by the semiconductor switches set on the path may increase. 
   In order to prevent this increase in heat generation, after a time during which both driving signal A and driving signal E are OFF, driving signal A and driving signal E are switched from OFF to ON. In this way, a short circuit preventing time (or dead time) is attached to the driving signal when generating pulses. 
   Just like the attachment of a dead time to driving signal A and driving signal E, a dead time is also attached to driving signal E and driving signal C. Also, in order to prevent a short circuit between the positive electrode and negative electrode, a dead time is attached to driving signal A and driving signal B, and to driving signal E and driving signal B. 
     FIG. 11  is a waveform diagram illustrating the generation of pulses of driving signal A and driving signal E by means of a sawtooth wave comparison. As shown therein, in order to attach a dead time to driving signal A and driving signal E in generating the driving signals, modulation rate instruction values mu_a up*, mu_a_c_down* offset by a dead time from modulation rate instruction value mu_a_c* are determined as follows:
   mu   —   a   —   c _up*= mu   —   a   —   c*+Hd ; and   mu   —   a   —   c _down*= mu   —   a   —   c*−Hd.    
Here, from the amplitude Htr (from bottom edge to apex) of the sawtooth wave, period Ttr, and dead time Td, Hd is determined as follows:
   Hd= 2 Td×Htr/Ttr.    
   The carrier is compared with the various modulation rate instruction values mu_a_c*, mu_a_c_up*, mu_a_c_down*, and the state of switching of driving signal A and driving signal E are determined according to the following rules: 
   1) If mu_a_c_down*≧carrier for DC power source  11   a , then driving signal A=ON; 
   2) If mu_a_c*≦carrier for DC power source  11   a , then driving signal A=OFF; 
   3) If mu_a_c*≧carrier for DC power source  11   a , then driving signal E=OFF; and 
   4) If mu_a_c_up*≦carrier for DC power source  11   a , then driving signal E=ON. 
   In this way, by generating the driving signal, it is possible to set dead time Td between driving signal A and driving signal E, and it is possible to prevent a short circuit between the positive electrodes. 
     FIG. 12  is a waveform diagram illustrating pulse generation of driving signal D and driving signal C by means of a triangular wave wave comparison. As shown therein, the pulse generation method when voltage pulses are output from DC power source  11   b  is the same as in the case of DC power source  11   a . In order to generate a driving signal with a dead time attached to driving signal D and driving signal C, modulation rate instruction values mu_b_c_up* and mu_b_c_down* offset by a dead time portion from modulation rate instruction value mu_b_c* are determined according to:
   mu   —   b   —   c _up*= mu   —   b   —   c*+Hd ; and   mu   —   b   —   c _down*= mu   —   b   —   c*−Hd.    
   These values are then compared with the carrier for DC power source  11   b . The state of the switch of driving signal D and driving signal C is determined according to the following rules: 
   1) If mu_b_c_down*≧carrier for DC power source  11   b , then driving signal D=OFF; 
   2) If mu_b_c*≦carrier for DC power source  11   b , then driving signal D=OFF; 
   3) If mu_b_c*≧carrier for DC power source  11   b , then driving signal C=OFF; and 
   4) If mu_b_c_up*≦carrier for DC power source  11   b , then driving signal C=ON. 
   In this way, it is also possible to set dead time Td between driving signal D and driving signal C, and it is possible to prevent a short circuit between the positive electrodes. 
     FIG. 13  illustrates the waveform of an example of generation of pulses with dead time Td attached. Driving signal E is a signal obtained by attaching dead time Td between it and driving signal A, and driving signal C is a signal obtained by attaching dead time Td between it C and driving signal D. Consequently, when driving signal B is generated as an AND function of driving signal E and driving signal C according to:
   B=E·C,    
it is possible to generate dead time Td in driving signal B and driving signal A and in driving signal B and driving signal E.
 
   Driving signal A through driving signal E for the various switches are input to driving signal processor part  31  shown in  FIG. 3 .  FIG. 14  is a circuit diagram illustrating the driving signal processor part  31 . An explanation is given here for signal processing for the U-phase. However, for phases other than the U-phase, the operation is the same as that of generation of signal of DC power source  11   a  shown in  FIG. 11 , and it will not be explained in detail again. 
   As shown in  FIG. 14 , driving signal processor part  31  has NOT circuits  47   a ,  47   b , AND circuits  48   a ,  48   b ,  48   c ,  48   d ,  48   e , NOR circuit  49  and OR circuits  50   a ,  50   b ,  50   c ,  50   d . Driving signal processor part  31  is a logic circuit that takes driving signals A-E of the various switches and ON/OFF judgment signal pwm_enable as inputs and outputs the driving signals of the various switches. Before performing a logic operation with the ON/OFF judgment signal pwm_enable, an AND operation is performed for driving signal B as logically inverted by NOT circuit  47   a  and driving signal A and driving signal D, respectively, by means of AND circuit  48   a  and AND circuit  48   b . Signals Ao, Do are respectively output. As a result, it is possible to prevent the output of a signal with simultaneous ON of driving signal A and driving signal B and simultaneous ON of driving signal D and driving signal B, and it is possible to prevent a short circuit between electrodes. 
   Also, for a L (Low) signal with both driving signal C and driving signal E OFF, the output of NOR circuit  49  becomes H (High). Since this signal and original driving signal E pass through OR circuit  50   a , signal Eo output from OR circuit  50   a  becomes H. Similarly, since driving signal C passes through OR circuit  50   b , signal Co output from OR circuit  50   b  becomes H. That is, when the signal passes to driving signal processor part  31 , when both driving signal E and driving signal C are OFF signals, the switches driven by driving signal E and driving signal C is turned ON, so that a current path from motor  15  to the power source can be guaranteed. 
   The AND values between ON/OFF judgment signal pwm_enable and signals Ao, Do, and signal Bo based on input driving signal B are respectively determined by AND circuits  48   c ,  48   d ,  48   e , and driving signals A, D, B are output. By means of this logic operation, when ON/OFF judgment signal pwm_enable is L, driving signals A, D, B become L, and the switches driven by these signals are OFF. Feeding of electric power from the power source to the load is stopped. 
   Also, the OR values between signals Co, Eo and the signal obtained by inverting ON/OFF judgment signal pwm_enable by NOT circuit  47   b  are determined by OR circuits  50   c ,  50   d , and driving signal E and driving signal C are output. Through this logic operation, ON/OFF judgment signal pwm_enable becomes L. That is, a stop signal for electric power converter  12  is received. Both driving signal C and driving signal E become H, and the switches driven by driving signal C and driving signal E are turned ON. 
   In this way, electric power converter  12  driven as described above allows operation of the electric power allotment of DC power source  11   a  and DC power source  11   b , and it is possible to perform allotment control for the power of conventional power converter. As a result, it is possible to realize a smaller size and higher efficiency for the overall electric power conversion system. 
   Also, by adding a voltage offset compensation value only to the phase where pulses are generated from plural power sources, the sum of the average ON times for one period of the electrical angle of semiconductor switch  23   a  (associated with signal A) and semiconductor switch  25   a  (associated with signal D) of the U-phase becomes different from the switches of the V-phase and W-phase (semiconductor switches  21   a ,  22   a ). Similarly, the average ON times of the switches (semiconductor switches  17   a ,  18   a ) connected to common negative electrode bus line  16  are different from each other. 
   By controlling so that these ON/OFF switch times are realized, even when the output voltage pulses are divided and allotted from DC power sources of different power source voltages, the average output voltages of the various phases in one period of the electrical angle are equal to each other. Hence, the motor can be driven with the same AC current waveform as that for conventional inverter driving, and it is possible to allot the power of power sources without generation of significant torque ripple in the motor current and without a significant decrease in the efficiency. 
     FIG. 15  is a circuit diagram illustrating another make up of the electric power converter shown in  FIG. 1 . As shown in  FIG. 15 , the switching device required to obtain a reverse blocking function is composed of a IGBT (Insulated Gate Bipolar Transistor) and diodes. Switches  25   a ,  25   b  are replaced with a serial connection of a group composed of IGBT  251   a  that stops current in the direction from the power source to the motor  15  and switches power feeding and diode  252   a  that allows flow of current only in the direction from the motor  15  to the power source, and a group composed of IGBT  251   b  that stops current from the motor  15  to the power source and switches power feeding and diode  252   b  that can stop current only in the direction from the power source to the motor  15 . 
   With this arrangement, it is possible to form switches  25   a ,  25   b  without using a reverse blocking-type IGBT. Similarly, the switching device formed by switches  23   a ,  23   b  can also be formed instead by a switching device including IGBT  231   a , diode  232   a , IGBT  231   b  and diode  232   b.    
     FIG. 16  is a circuit diagram illustrating yet another arrangement of the electric power converter shown in  FIG. 1 . In this arrangement, among the two power sources, one power source Vdc_b is a 42V type alternator  110   b , and the other power source Vdc_a is a 14V type battery  110   a . For a vehicle having this 2-power source system carried on it, in order to eliminate the instability of the voltage generated by the 42V type alternator  110   b , usually connected to the engine or another primary driving machine, it is sometimes desirable to set another 42V type battery. 
   Also, in a vehicle, 14V type battery  110   a  can be present as a battery for auxiliary equipment for turning ON the head lights, etc. In this case, when electric power is charged in 14V type battery  110   a , the voltage of 42V type alternator  110   b  is lowered by a DC/DC converter or the like for feeding. 
   In this arrangement, it is possible to perform the same function as that of a combination of a single 42V type alternator and a single 14V type battery. The mechanism is as follows. For the 14V type battery  10   a , a portion of the electric power fed from 42V type alternator  110   b  to motor  15  is fed back via switch  25   b  to 14V type battery  110   a  so that 14V type battery  110   a  can be charged. When the voltage of the power generated by 42V type power generator  110   a  is unstable, the electric power of 14V type battery  110   b  is fed through switch  25   a  to the motor  15 , so that the instability in driving of the motor  15  can be alleviated. 
   As explained above, in the present embodiment, the scheme is not limited to plural DC power sources. It is also possible to use various combinations, such as a combination of a fuel cell and a battery, a combination of a battery and another battery, a combination of an alternator and a battery, etc. The alternator may also be replaced by an AC current source via a rectifier, such as so-called commercial AC power. 
   In the following, an explanation is given regarding the effects of the present embodiment. 
   The arrangement of the embodiment is as follows. Plural power sources are DC power sources of different potentials. For the switching devices of the phase connected to the plural power sources, the switches connected to the lowest potential are active elements without a reverse blocking function and diodes, and the remaining are made of elements having a reverse blocking function. The phase connected to one DC power source has all of its arms composed of active elements without a reverse blocking and diodes. As a result, it is possible to reduce the number of elements. 
   Also, the switch of the phase with the output connected to the plural power sources generates pulses with the average value of the ON time for one period of the electric angle different from the average value of the ON time for one period of the electric angle of the switch of the phase connected to one power source. Consequently, even when AC voltage is output from power sources with different voltages, by generating pulses with different ON time average values it is possible to output an average voltage value just like the AC voltage output from the phase connected to one power source. By identical setting of the level of the average voltage value, no offset current flows in the AC current, and it is possible to operate the motor without torque ripple and without a decrease in the efficiency of the motor. 
   The phase voltage instruction value of the phase that generates pulses from the plural power sources and synthesizes them for output has an offset value with respect to the phase voltage instruction value of the other phase. Consequently, even if the output voltage pulses are divided and allotted from DC power sources having different power source voltages, it is still possible to have the same average output voltage for the various phases in one period of the electrical angle, and it is possible to drive with a motor current just like that in conventional inverter driving. It is further possible to allot the power of the power sources without generation of significant torque ripples in the motor current and without a decrease in efficiency. 
   In addition, the phase voltage instruction value of the phase that generates pulses from the plural power sources and synthesizes them for output is obtained by adding or subtracting the offset value with respect to the phase voltage instruction value of the other phase to/from the phase voltage instruction value before allotment of each power source. Consequently, it is possible to allot the power of the power sources without generation of significant torque ripple in the motor current and without a decrease in efficiency. Also, without changing control of the other phase from the well-known method of control of an inverter, one may adopt an embodiment of the control method taught herein for controlling only the phase that outputs pulses from plural power sources. As a result, one may adopt an arrangement simply by addition to control an electric power converter of the prior art. 
   Also, the offset value is computed from the voltage values of the power sources and the proportions of allotment of the phase voltage instruction value to the power sources. Accordingly, without newly detecting the output error voltage, it is possible to use information stored in the controller to allot the power of the power sources without generation of significant torque ripple in the motor current and without a decrease in efficiency. 
   In the following, an explanation is given regarding an electric power conversion control system as a second embodiment of the invention. An explanation is given only for differences from the first embodiment. For the electric power converter of the electric power conversion control system of the second embodiment, it is possible to perform pulse generation from plural power sources only for the U-phase. On the other hand, the electric power converter of the electric power conversion control system according to the second embodiment has a circuit arrangement that can perform pulse generation also in the V-phase. 
     FIG. 17  is a block diagram illustrating the electric power conversion control system  55  according to the second embodiment. As shown in  FIG. 17 , electric power conversion control system  55  outputs modulation rate instruction value mv_b_c* from electric power controlling/modulating arithmetic operation part  29  of electric power controller  14   a  to PWM pulse generator  30 . Also, the system has an electric power converter  56  with an arrangement different from that of electric power converter  12 . The other features are the same as those of electric power conversion control system  10  in the first embodiment. 
     FIG. 18  is a circuit diagram illustrating details of the electric power converter  56  shown in  FIG. 17 . Positive electrode bus line  20  of DC power source  11   a  and the V-phase terminal of motor  15  are connected to each other via a group of semiconductor switches  21   a ,  57 . Positive electrode bus line  24  of DC power source  11   b  and the V-phase terminal of motor  15  are connected to each other via two semiconductor switches  58   a ,  58   b  that allow control of bidirectional conduction. 
   That is, between positive electrode bus line  20  and the V-phase terminal of motor  15 , in place of diode  21   b , semiconductor switch  57  is set. Between positive electrode bus line  24  and the V-phase terminal of motor  15 , a group of two semiconductor switches  58   a ,  58   b  are newly set. The remaining features of the arrangement and operation are the same as those in electric power converter  12  as shown in  FIG. 1 . 
   In this second embodiment, the arithmetic operation of the modulation rate of the V-phase is not performed using multiplier  40  as shown in  FIG. 6 , and the operation is performed by way of an electric power control/modulation rate arithmetic operation part  29  with the same arrangement as that of electric power controlling/modulating arithmetic operation part  29  that performs the electric power control/modulation rate arithmetic operation of the U-phase according to  FIG. 5 . Then, the voltage offset compensation operation is also performed in the same way for the U-phase. 
   In the following, an explanation is given regarding the electric power conversion control system in a third embodiment of the invention. An explanation is given only for the differences between it and the first embodiment. In this third embodiment, the arithmetic operation in determining the voltage offset compensation value of the electric power converter is partially different from that of the electric power converter  12  in the first embodiment. 
     FIG. 19  is a block diagram illustrating the modulation rate offset arithmetic operation part  59  in the third embodiment. Voltage offset arithmetic operation part  59  has voltage offset compensation arithmetic operation unit  60 , dq/3-phase conversion part  61 , subtractor  62 , U-phase current control part  63  and adder  64 . 
   Voltage offset compensation arithmetic operation unit  60  performs the same arithmetic operation as that of voltage offset compensation arithmetic operation unit  46  in the first embodiment as shown in  FIG. 8 , and it outputs feedforward voltage offset compensation value v_o_ff*. In addition, with feedback control of the phase current, U-phase current control part  63  computes feedback voltage offset compensation value v_o_fb*. Here, from d-axis current instruction value id* and q-axis current instruction value iq* as well as phase θ, coordinate transformation is performed by dq/3-phase conversion part  61 . Then, phase current instruction value iu* of the U-phase is determined. Using subtractor  62 , the difference from U-phase phase current iu detected by a current sensor is computed, and, by means of proportional-integration (P-I) control, U-phase current control is performed to determine v_o_fb*. Then, using adder  64 , voltage offset compensation values v_o_ff* and v_o_fb* are added to determine modulation rate offset value v_o*. 
   In this way, by computing the offset value by means of current feedback control while the offset value is computed from the voltage of the power source and the offset in the modulation rate, even if there is an undetermined external disturbance voltage in addition to a difference in the output voltage of Δvu_out_ave due to the ON resistance of the switch and the ON/OFF time delay or the like, the offset portion of the voltage is compensated by the feedback control of the current. The output voltage average value of the U-phase becomes equal to that of the other phase. Consequently, it is possible to suppress the offset of the current in each phase, and it is possible to allot the power of the power source without generation of torque ripple and without a decrease in efficiency. 
     FIG. 20   a  is a waveform diagram illustrating the case when voltage offset compensation is not performed by means of current feedback control, and  FIG. 20   b  is a waveform diagram illustrating the case when addition is performed for the voltage offset compensation value in this embodiment. In the waveform of the phase current before addition of the voltage offset compensation value, in company with a change in electric power distribution proportion rto_pa, an offset current appears in the waveform of the phase current as shown in  FIG. 20   a . In the waveform of the phase current when addition is performed for the voltage offset compensation value in this embodiment, no offset current appears as shown in  FIG. 20   b , and suppression of current offset can be seen. 
   Next, an explanation is given regarding the effect of the third embodiment. This embodiment has a portion that detects the phase current of the phase that generates pulses from the plural power sources and synthesizes them for output. Also, feedback control is computed from the difference between the phase current instruction value and phase current of that phase. The output of this feedback control is taken as the offset value. As a result, without newly detecting the output error voltage, it is possible to allot the power of the power sources by means of information stored in the controller without generation of torque ripple and without a decrease in efficiency. 
   Also, this embodiment has a portion that detects the phase current of the phase that generates pulses from the plural power sources and synthesizes them for output. Another portion extracts the DC current component of that phase current from the detection value of that phase current. Feedback control is computed from the difference between the DC current instruction value and the DC current component of that phase current. The output of this feedback control is taken as the offset value. Consequently, without newly detecting the output error voltage, it is possible to allot the power of the power sources by means of information stored in the controller without generation of torque ripple and without a decrease in efficiency. 
   Also, this embodiment has a portion that detects the phase current of the phase that generates pulses from the plural power sources and synthesizes them for output. This embodiment also computes the offset value as the sum of the offset value as the output of the arithmetic operation from the difference between the phase current instruction value and the phase current of the phase that generates pulses from the plural power sources by means of feedback control and the offset value computed from the power source voltages of the plural power sources, the average value of the pulse width instruction values of the power sources and the average value of the pulse width instruction values of the other phase. 
   As a result, without newly detecting the output error voltage, it is possible to allot the power of the power sources by way of information stored in the controller without generation of torque ripple and without a decrease in efficiency. Also, because current feedback control is performed for the offset current that cannot be compensated only by the offset value computed from the voltage value and the allotment proportion, this is preferable due to further reduction in the offset current. Also, compared with the case of feedback control of current alone, it is possible to improve the response property in suppressing the offset current. 
   An explanation is next given regarding the electric power conversion control system in the fourth embodiment of the invention. The electric power converter of the control system in the fourth embodiment is different from the electric power converter in the third embodiment with respect to the portion from the feedback control of the phase current to compute v_o_fb*. 
     FIG. 21  is a block diagram illustrating the modulation rate offset arithmetic operation part  65  in the fourth embodiment. Modulation rate offset arithmetic operation part  65  has voltage offset compensation arithmetic operation unit  60 , low-pass filter (LPF)  66 , subtractor  62 , U-phase current control part  63  and adder  64 . 
   Voltage offset compensation arithmetic operation unit  60  outputs voltage offset compensation value v_o_ff*. LPF  66  lets U-phase current iu detected by the current sensor pass through it, and outputs current value iu 0 . Current value iu 0  becomes the value obtained by extraction of the DC current component contained in U-phase current iu. After determining the difference between current value iu 0  and instruction value iu 0 * of the DC current component of the U-phase using subtractor  62 , U-phase current control consisting of proportional-integration (P-I) control is performed to determine feedback voltage offset compensation value v_o_fb* using U-phase current control part  63 . Here, instruction value iu 0 *=0. Then, feedforward voltage offset compensation value v_o_ff* and feedback voltage offset compensation value v_o_fb* are added using adder  64  to determine modulation rate offset value v_o*. 
   In this way, by extracting the DC current component contained in the phase current, and controlling to 0 with feedback control, the offset current contained in the phase current can be controlled at nearly 0. As a result, it is possible to allot the power of the power sources without generation of torque ripple and without a decrease in efficiency. 
   In the following, an explanation is given regarding the electric power conversion control system in the fifth embodiment of the invention. The electric power converter in the fifth embodiment differs from the electric power converter in the first embodiment with respect to the arrangement of the electric power control/modulation rate arithmetic operation part. 
     FIG. 22  is a block diagram illustrating the electric power control/modulation rate arithmetic operation part  67  in the fifth embodiment. Electric power control/modulation rate arithmetic operation part  67  has multiplier  37 , subtracters  36 ,  68 ,  69 , modulation rate arithmetic operation part  38 , modulation rate amendment (or correction) part  39  and switch  70 . 
   Using switch  70 , selection is made between adding computed voltage offset compensation value v_ 0 * to DC power source  11   a  or to DC power source  11   b . That is, after the output from switch  70  is input to subtracter  68  or subtracter  69  for arithmetic operation, voltage instruction value vu_a* on the side of DC power source  11   a  or voltage instruction value vu_b* on the side of DC power source  11   b  is input to modulation rate arithmetic operation part  38 . Switching by switch  70  is performed with a switch selecting signal that selects the power source having the higher voltage between DC power source  11   a  and DC power source  11   b.    
   By adding compensation voltage on the side of the power source with tolerance in the output voltage, it is possible to realize a voltage instruction by using the power source having a higher voltage with tolerance in the output of voltage pulses corresponding to the offset value. For example, the case of a power source with tolerance among plural power sources can be coped with according to the residual quantity of the electric power, and it is possible to perform compensation control in a wide operating range. 
   Next, an explanation is given regarding the effects of the fifth embodiment. 
   In the fifth embodiment, the phase voltage instruction value of the phase that generates pulses from the plural power sources and synthesizes them for output is obtained by adding or subtracting the offset value to/from any of the phase voltage instruction values allotted to any of the power sources. Consequently, it is possible to realize voltage instruction using a power source with tolerance in output of the voltage pulses corresponding to the offset value among several power sources. 
   Also, the phase voltage instruction value of the phase that generates pulses from the plural power sources and synthesizes them for output is obtained by adding or subtracting the offset value to/from a phase voltage instruction value generated from the power source having the highest power source voltage among the power sources. Because of this, it is possible to realize a voltage instruction using a power source with tolerance in output of the voltage pulses corresponding to the offset value among several power sources. 
   The electric power conversion control system in a sixth embodiment of the invention is next discussed. The arrangement of the torque control part  77  in the electric power conversion control system  75  in the sixth embodiment is different from the electric power conversion control systems according to the other described embodiments. 
     FIG. 23  is a block diagram illustrating the arrangement of the electric power conversion control system  75  in the sixth embodiment. Electric power conversion control system  75  has torque control part  77  in electric power controller  76  instead of torque controller  13 . The remaining features of the arrangement and operation are the same as those of electric power conversion control system  10  in the first embodiment. 
   Torque control part  77  computes d-axis current instruction value id* and q-axis current instruction value iq* of motor  15  from torque instruction value Te* and motor rotation velocity ω applied from the outside, as well as electric power instruction value Pa* of DC power source  11   a , electric power instruction value Pb* of DC power source  11   b  and electrical angle θ of motor  15 . A block diagram illustrating the details of the torque control part  77  is shown in  FIG. 24 . 
   As shown therein, torque control part  77  has torque controller  78 , charge power controller  79 , control mode switch  80  and current instruction value switch  81 . Torque controller  78  takes as reference a map that has Te* and ω as axes and is prepared beforehand to output id 1 *, iq 1 *. charge power controller  79  takes electric power instruction values Pa*, Pb* and electrical angle θ as inputs, and outputs id 2 * and iq 2 *. At the same time, charge power controller  79  outputs electrical angle θ′ and electric power distribution target value rto_pa of DC power source  11   a.    
   Also, when both the magnitude of the torque and rotation velocity c are near 0, the control mode switch  80  selects id 2 *, iq 2 * output from charge power controller  79 . Otherwise, outputs id 1 *, iq 1 * from torque controller  78  are selected. 
   In charge power controller  79 , from electric power instruction value Pa* of DC power source  11   a  and electric power instruction value Pb* of DC power source  11   b , electric power distribution target value rto_pa is first computed using the following formula:
 
 rto   —   pa=Pa */( Pa*+Pb* )
 
     FIG. 25  is a flow chart illustrating the processing flow of the electric power controller  76  in the sixth embodiment. When the current instruction value and electrical angle θ′ are generated, |cos θ| computed from electrical angle θ of motor  15 , which angle is obtained from the position sensor of the motor, is compared with prescribed value TH 0  to judge whether |cos θ| is larger than TH 0 . That is, in step S 101 , a query is formed as |cos θ|&gt;TH 0 . The value of TH 0  is explained later. When cos θ is equal to 0 or near 0, the d-axis current flowing in the U-phase is very small, so it is difficult to control charging using only the d-axis current, and this value is set for judging whether it is possible to control charging by means of the d-axis current alone. 
   If the judgment result, i.e., the response to the query |cos θ|&gt;TH 0  is yes, the sign of cos θ is judged in step S 102 . When cos θ is positive in response to the query cost θ&gt;0?, the sign of id 2 * is set negative in step S 103 . If cos θ is not positive in step S 102 , the sign of id 2 * is set positive in step S 104 . Next, in step S 105 , in order to ensure that motor  15  does not generate a torque, iq 2 * is set as iq 2 *=0. Also, id 2 * is generated with reference to a map based on electric power instruction value Pb* of DC power source  11   b . This map is a one-dimensional map that outputs the magnitude of id 2 *, and it is prepared experimentally beforehand and is stored inside charge power controller  79 . 
   Current instruction value id 2 * is next generated in step S 106  from the sign information and the size of id 2 *. Then, as electrical angle θ′ for use in coordinate transformation of the d·q axis current control, the electrical angle θ obtained in step S 107  by the position sensor of motor  15  is substituted (θ′=θ) for use. Processing is then stopped. 
   On the other hand, if instead |cos θ| is less than or equal to TH 0  in response to the query of step S 101  (NO), id 2 * and iq 2 * are generated with reference to a map in step S 108  based on electric power instruction value Pb* of DC power source  11   b . This map is also prepared experimentally beforehand and stored in electric power controller  79 . Here, any values may be used as id 2 * and iq 2 *. As to be explained later, by using the virtual electrical angle from id 2 * and iq 2 *, voltage instruction value vu* of the U-phase, voltage instruction value vv* of the V-phase and voltage instruction value vw* of the W-phase are generated such that an AC current that vibrates at high frequency flows. On the other hand, by means of the electric power allotment control explained above, feeding/charging of each power source is compensated in a feedback way. As a result, charging can be performed for any values set at id 2 *, iq 2 *. In order to facilitate explanation, just like the case where |cos θ|&gt;TH 0 , id 2 * is generated from the map based on electric power instruction value Pb*, and iq 2 * is set at 0. 
   In next step S 109 , virtual electrical angle θ″ is computed. Virtual electrical angle θ″ has a value that is computed successively based on the frequency of the virtual electrical angle. The virtual electrical angle frequency is set in the range of several hundred Hz to several kHz. Then, in step  110 , as electrical angle θ′ for use in coordinate transformation of the d·q axis current control, virtual electrical angle θ″ is substituted according to θ′=θ″. Then processing comes to an end. In this case, instead of control of the d·q axis current commonly known for the motor  15 , current control is performed in a rotating coordinate system that undergoes virtual rotation. 
   Next, an explanation is given regarding the background of this control. First, the following relationship exists between the d·q axis current and the 3-phase AC current: 
   
     
       
         
           
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   For electric power converter  12  as shown in  FIG. 1 , the circuit arrangement is such that the U-phase voltage is generated from two DC power sources  11   a ,  11   b . It is possible to feed the electric power of DC power source  11   b  to motor  15  or to charge electric power only via the circuit of the U-phase. While motor  15  is stopped near velocity  0 , as DC power source  11   b  is charged without generating torque for motor  15 , control may be performed such that iq=0. 
   Here, with iq=0, in consideration of current iu of the U-phase, one has:
 
iu=id cos θ.
 
That is, it can be seen that iu is determined by cos θ determined from d-axis current id and the position of the rotor. Regarding the absolute value of id, |id|, it is possible to select a prescribed value based on the electric power allotment ratio. On the other hand, in order to avoid influencing the torque by the sign of the id, it is possible to select any sign. Here, the sign is determined in consideration of the movement direction of the electric power. That is, in the direction of charging of DC power source  11   b , current may flow in the negative direction of the U-phase current such that current flows in the direction from motor  15  to electric power converter  12 . The sign of id is selected to ensure that the sign of iu is negative. Consequently, if cos θ is positive, id is selected negative. If cos θ is negative, id is selected positive. On the other hand, when DC power source  11   a  is charged, the sign of id may be selected such that the sign of iu becomes negative or the sign of iu becomes positive. Selection of sign is valid in the case when the charge/discharge phase is limited as in this embodiment.
 
   On the other hand, judging from this formula, when cos θ is 0, no matter how id is selected it is not possible to manipulate the electric power of DC power source  11   b  via the U-phase. When cos θ is 0 or is near 0, iq, id are taken as high frequency current, and the motor current is controlled while the average torque of motor  15  is set at 0. If id, iq have a sufficiently high frequency, the average torque in a short time becomes 0 although an instantaneous torque is generated in motor  15  so that motor  15  does not rotate in one direction. 
   That is, by giving a virtual electrical angle of vibration at a high frequency as mentioned, it is possible for the average torque to be 0, and current flows in the U-phase. 
     FIG. 26  schematically illustrates selection of the sign of the d-axis current and selection of the high frequency current. As shown therein, selection of the sign of d-axis current id and selection of the high frequency current are performed as described above. 
     FIG. 27  illustrates an example of the results of electric power control in the sixth embodiment. As shown, motor  15  is stopped, and the electric power instruction value is set such that DC power source  11   a  (power source a) feeds electricity while DC power source  11   b  (power source b) is charged. Accordingly, even if motor  15  is stopped at various electrical angles, it is still possible to control the electric power so that charging is performed from one of DC power source  11   a  and DC power source  11   b  to the other. 
   In this embodiment, the U-phase is the phase that can perform pulse generation from plural power sources. But in the V-phase and W-phase, when the currents of the various phases are selected such that current flows on the path connected to DC power source  11   b , it is possible to perform the same electric power control as this embodiment. Also, for all of the phases, it is possible to control the electric power also by flowing current at the virtual electrical angle frequency. As also shown in this embodiment, by switching the virtual electrical angle frequency and the d-axis current, it is possible to reduce the probability of generation of the influence of vibration, magnetic noise, etc., if they become a problem due to variation in the torque of the motor caused by the virtual electrical angle frequency current. 
   In the following, an explanation is given regarding the effects of the sixth embodiment. 
   This embodiment has a part that generates a current instruction value for the multi-phase AC motor and the allotment proportions for the power source voltage from the electrical angle, rotation velocity and electric power instruction values of the power sources of the multi-phase AC motor. The output voltage is generated based on the current instruction value for the multi-phase AC motor, and the output voltage is allotted to the power sources. Consequently, even if the prescribed phase has a circuit arrangement wherein pulses are generated from plural power sources and are synthesized for output, because the power feeding current instruction value is generated from the electric power instruction value and the electrical angle and rotation velocity of the motor, no matter what the motor position and velocity are, the power of plural power sources is allotted to the power sources, and it is possible to control the electric power. As a result, even if the motor is stopped, by generation and synthesis of pulses from plural power sources, allotment to plural power sources and charging from one power source to another power source without using a DC/DC converter or the like are still possible. 
   Also, in this embodiment the part that generates the current instruction value for the multi-phase AC motor and the allotment proportions for the power of the power sources generates the magnitudes of the d-axis current instruction value and the q-axis current instruction value. Together with these values, this embodiment switches the sign of the d-axis current instruction value based on the value of the electrical angle of the multi-phase AC motor. Consequently, by generating the instruction value of the d·q axis current and switching the sign of the d-axis current, it is possible to allot the power for plural power sources to each of the power sources and to control the electric power independent of the position of the motor. 
   In the sixth embodiment, the sign of the d-axis current instruction value is such that the current path connected to the power source that charges the electric power is selected in the direction of flow of the charging current based on the electric power instruction value for the plural power sources. By selecting the switching of the sign of the d-axis current such that current flows in the path for charging electric power, it is possible to allot the power for plural power sources to each of the power sources and to control the charging electric power of the power sources. 
   Also, this embodiment has a part that generates the virtual electrical angle of the multi-phase AC motor. The part that generates the current instruction value for the multi-phase AC motor and the allotment proportions for the power of the power sources generates the current instruction value at the virtual electrical angle frequency. By generating the virtual electrical angle of the motor and generating the current instruction value of the virtual electrical angle frequency, current at the frequency of the virtual electrical angle can flow to the motor. As a result, no matter where the rotor of the motor stops, it is possible to allot the power for plural power sources to each of the power sources and to control the charging electric power for the power sources by means of current at the virtual electrical angle frequency. 
   This sixth embodiment has a part that generates the voltage instruction value in a rotating coordinate system that rotates at the virtual electrical angle based on the motor current instruction value in the rotating coordinate system. The output voltage instruction value is generated by means of coordinate transformation of the voltage instruction value in the rotating coordinate system using the virtual electrical angle in the stationary coordinate system. Consequently, by generating the voltage instruction value in the rotating coordinate system based on the motor current instruction value in the coordinate system rotating at the virtual electrical angle, and by generating the output voltage instruction value by means of coordinate transformation using the virtual electrical angle, even if the frequency of the virtual electrical angle is high, highly precise control of the current at high frequency is possible just as in conventional vector control. Highly precise control of the charging electric power for the power sources is also possible. Further, the motor is an AC motor. If current control is performed using vector control, it is possible to use the controller as is to control the rotating coordinates at the virtual electrical angle, and it is possible to eliminate an increase in the cost of the controller since the number of parts to be added to the controller is decreased. 
   Also, based on the electrical angle of the multi-phase AC motor, the current instruction value is selected from the magnitudes of the d-axis current instruction value and the q-axis current instruction value, the current instruction value due to switching of the sign of the d-axis current instruction value and the current instruction value at the virtual electrical angle frequency. Consequently, it is possible to allot the power for plural power sources to each of the power sources with only the d-axis and q-axis current, and it is also possible to control the charging electric power for the power sources. In addition, no matter where the motor rotor stops, control of the electric power is still possible by using the current instruction value of the virtual electrical angle frequency. 
   Next, an explanation is given regarding the electric power conversion control system in the seventh embodiment of the invention. Electric power conversion control system  82  of the seventh embodiment differs from the sixth embodiment that performs pulse generation from plural power sources only in the U-phase in that it uses electric power converter  56  having a circuit arrangement that allows generation of pulses from plural power sources also in the V-phase as described with reference to  FIG. 21 . The seventh embodiment also has a torque control part  84  with an arrangement different from the torque control part in the sixth embodiment. The other features of the arrangement and operation are the same as those of electric power conversion control system  75  in the sixth embodiment as shown in  FIG. 23 . 
     FIG. 28  is a block diagram illustrating the arrangement of the electric power conversion control system  82  in the seventh embodiment. As shown in  FIG. 28 , torque control part  84  in electric power controller  83  is different from torque control part  77  of  FIG. 23 . Also, electric power conversion control system  82  has electric power converter  56 . 
     FIG. 29  is a block diagram illustrating in detail the arrangement of the torque control part shown in  FIG. 28 . Instead of charge power controller  79  previously taught, torque control part  84  has charge power controller  85  without the output of electrical angle θ′. Otherwise, torque control part  84  has the same arrangement as that of torque control part  77  previously described. 
     FIG. 30  is a flow chart illustrating the processing flow of the electric power controller in the seventh embodiment. First, judgment is made as to whether electrical angle θ of motor  15  is smaller than prescribed phase θth 2  that is larger than prescribed phase θth 1  in step S 201 . The value of electrical angle θ of motor  15  is obtained from the position sensor of motor  15 , and this electrical angle θ is compared with prescribed phases θth 1  and θth 2 . By example, θth 1 =−π/6 and θth 2 =5π/6. 
   As the result of this judgment, if electrical angle θ enters this range (i.e., the response to the query of step S 201  is YES), the sign of id 2 * is taken as negative in step S 202 , and, if electrical angle θ is out of this range (i.e., the response to the query of step S 201  is NO), the sign of id 2 * is taken as positive in step S 203 . Then, in step S 204 , iq 2 * is set at 0 such that motor  15  generates no torque, and id 2 * is generated based on electric power instruction value Pb* of DC power source  11   b  with reference to a one-dimensional map that outputs the magnitude of id 2 *. This map is prepared experimentally and is stored beforehand inside the device. 
   Then, from the sign information of id 2 * and its magnitude |id 2 *|, current instruction value id 2 * is generated in step S 205 . Processing comes to an end. 
   In this way, by selecting the sign of id 2 *, current flows from motor  15  to the power source in one of the U-phase and V-phase so that it is possible to charge DC power source  11   b . In this way, even if the motor is stopped, allotment control of the electric power including charging of DC power source  11   b  is still possible. 
   In this embodiment, the U-phase and V-phase are set as the phases that can generate pulses from plural power sources. However, it is also possible to use a combination of the two remaining phases of the three phases. Prescribed phases θth 1 , θth 2  can be selected to match the phases. Also, when charging is performed from DC power source  11   b  to DC power source  11   a , reverse selection of the sign of id* is possible. 
   In the following, an explanation is given regarding the electric power conversion control system in the eighth embodiment of the invention. The electric power conversion control system in the eighth embodiment has a braking device (brake) on the output shaft of the motor  15 .  FIG. 31  illustrates the braking device installed on the output shaft of the motor  15 . Load R is connected to output shaft  15   a  of motor  15 , and braking device  86  for mechanically braking motor  15  is installed at an intermediate point of output shaft  15   a.    
   A block diagram shown in  FIG. 32  illustrates in detail the torque control part  86  in the eighth embodiment. As shown therein, instead of charge power controller  79 , braking device  86  has charge power controller  87 . This charge power controller  87  outputs a braking device operating signal as the operating signal of braking device  86  in place of electrical angle θ′ output from charge power controller  79 . 
     FIG. 33  is a block diagram illustrating details of the charge power controller  87  shown in  FIG. 32 . Charge power controller  87  has iu map part  88 , iv, iw arithmetic operation part  89  and coordinate converting part  90 . Using iu map part  88 , and taking the map based on electric power instruction value Pb* as a reference, U-phase current instruction value iu* is determined. Using determined U-phase current instruction value iu*, a negative value map is formed to allow charging of DC power source  11   b.    
   The phase current instruction values of the other phases, that is, V-phase current instruction value iv* and W-phase current instruction value iw* are computed using the following formulas in iv, iw arithmetic operation part  89  so that 3-phase equilibrium is realized:
 
 iv*=−iu*/d ; and
 
 iw*=−iu*/ 2.
 
   From current instruction values iu*, iv*, iw* of the various phases and electrical angle θ, current instruction values id 2 *, iq 2 * of the d·q axis current are computed using coordinate converting part  90 . 
   Also, under the condition that d·q axis current instruction values id 2 * and iq 2 * are selected using control mode switch  80 , the operating signal of the braking device output from charge power controller  87  turns braking device  860 N. As a result, even at a position of electrical angle θ such that torque is generated by motor  15 , it is possible to keep output shaft  15   a  stopped without rotation. As a result, it is possible to control charging from one power source to another power source while motor  15  is held as stopped without generating variation in the torque of the motor. 
   In the following, an explanation is given regarding the electric power conversion control system in a ninth embodiment of the invention. This embodiment has a clutch device in place of the braking device on the output shaft  15   a  of the motor  15 . The remaining features of this embodiment are the same as those in the electric power conversion control system in the eighth embodiment. 
     FIG. 34  illustrates the clutch device installed on the output shaft of the motor in the ninth embodiment. Load R is connected to output shaft  15   a  of motor  15  of the electric power conversion control system, and clutch device  91  for mechanically releasing motor  15  and the load shaft is installed at an intermediate point of output shaft  15   a.    
     FIG. 35  is a block diagram illustrating in detail the torque control part  92  in the ninth embodiment. Instead of charge power controller  87 , torque control part  92  has charge power controller  93 . This charge power controller  93  outputs a clutch operating signal as the operating signal of clutch device  91  in place of the braking device operating signal output from charge power controller  87 . 
   Charge power controller  93  receives electric power instruction value Pb* and outputs id 2 *, iq 2 * using a map as a reference. Then, just as in the processing of charge power controller  79  in the sixth embodiment (see  FIG. 25 ), |cos θ| computed from electrical angle θ of motor  15  is compared with the prescribed value TH 0  to judge whether |cos θ| is larger than TH 0 . 
   In response to this query, if |cos θ|&gt;TH 0 , then iq 2 * is set to 0, and the sign of id 2 * is judged. On the other hand, if the relationship |cos θ|&gt;TH 0  is not met, the map is taken as a reference, and iq 2 * is output. At the same time, a clutch operating signal is output, and load R and output shaft  15   a  of motor  15  are released. Depending on the output of iq 2 *, a motor torque is generated in rotation. As a result, by controlling the d-axis current, charging control of DC power source  11   a  and DC power source  11   b  using an electric power allotment target value is possible. 
     FIG. 36  is a circuit diagram illustrating the electric power converter in a tenth embodiment of the invention. Electric power converter  96  of the tenth embodiment differs from that as shown in  FIG. 36 . Electric power converter  96  has positive electrode bus  20  of DC power source  11   a  and the W-phase terminal of motor  15  connected via a group of semiconductor switches  22   a ,  94 , and it has positive electrode bus  24  of DC power source  11   b  and the W-phase terminal of motor  15  connected via a group of two semiconductor switches  95   a ,  95   b  that allow control of bidirectional conduction. 
   That is, between positive electrode bus  20  and the W-phase terminal of motor  15 , semiconductor switch  94  is set in place of diode  21   b . Between positive electrode bus  24  and the V-phase terminal of motor  15 , a group of two semiconductor switches  95   a ,  95   b  is newly set. The remaining arrangement and operation are the same as those of electric power converter  56  (see  FIG. 18 ). 
   Although the arrangement of this circuitry is previously-known, as shown in  FIG. 37 , electric power controller  99  has electric power control/modulation rate arithmetic operation part  100 , which results in a different control method. Electric power control/modulation rate arithmetic operation part  100  has the following modes. First, an A-mode wherein plural power sources work in the U-phase, V-phase and W-phase in a like manner to that previously-known, a B-mode wherein plural power sources work only in the U-phase just like the electric power control/modulation rate arithmetic operation part in the first embodiment, a C-mode wherein the U-phase in the first embodiment is changed to the V-phase, a D-mode where the U-phase of the first embodiment is changed to the W-phase, an E-mode wherein plural power sources work only in the U-phase and V-phase like the electric power control/modulation rate arithmetic operation part in the second embodiment, an F-mode wherein the U-phase and V-phase of the second embodiment are changed to the W-phase and V-phase and a G-mode wherein the U-phase and V-phase of the second embodiment are changed to the U-phase and W-phase. In B-mode through G-mode, there is a mode for the exchange of power source a and power source b. 
   Since the mode is switched by mode switching part  100 , electric power control is possible without using a switch. That is, when a certain switch cannot be used due to a switch problem or the like, in the known art the corresponding power source cannot be used. However, in this tenth embodiment, it is possible to continue electric power control by switching the mode matched to the switch that is out of order. 
   In the tenth embodiment, even in the case of a defective switch or the like, it is possible to continue electric power control by means of a combination of the electric power controller in the first and second embodiments. For example, when a fuel cell is used in power source b, in the previously-known design, electric power control has to be continued with only the electric power left in power source a when a switch on the side of power source b is out of order. However, in the tenth embodiment, use of the residual phase to continue the electric power control with continued charging of power source a is preferable. 
   As explained above, according to the embodiments of the invention, the electric power converter that drives a multi-phase AC motor has a phase in which a driving voltage for driving the multi-phase AC motor connected to plural power sources is generated by generating and synthesizing output voltages of the plural power sources. Also, a phase in which the driving voltage of the multi-phase AC motor connected to one DC power source is generated by generating pulses from the output voltage of the DC power source. As a result, use and allotment of the power of plural power sources with fewer semiconductor elements are possible. 
   In the above, an explanation is given for the invention with reference to figures and application examples. However, those skilled in the art can make various modifications and amendments based on the present disclosure. Consequently, one should understand that those modifications and amendments are also included in the range of the invention. For example, the power sources that can be used in embodiments of the invention are not restricted to DC power sources. At least of one of the power sources may be any DC battery and capacitor that allows charge/discharge. Other power sources include fuel cells, uni-phase alternators, multi-phase alternators rectified by an inverter or the like, uni-phase AC commercial power sources, etc. 
   For example, as shown in  FIG. 38 , that connection can be made via a rectifier to avoid inversion of the voltage when using a uni-phase AC power source. Especially when a uni-phase AC power source  120   b  is used to control charging as described in the sixth through ninth embodiments, the scheme shown in  FIG. 38  is useful. It is possible to adopt a simple construction for the charger in the equipment used by charging, such as electric automobiles that can be charged to run using a commercial power source. 
   Also, the above-described embodiments have been described in order to allow easy understanding of the present invention and do not limit the present invention. On the contrary, the invention is intended to cover various modifications and equivalent arrangements included within the scope of the appended claims, which scope is to be accorded the broadest interpretation so as to encompass all such modifications and equivalent structure as is permitted under the law.