Patent Publication Number: US-2005116857-A1

Title: Method and dual-frequency gps receiver

Description:
The invention relates to satellite radionavigation, in particular satellite radionavigation of GPS (Global Positioning System), Galileo, GLONASS (Global Navigation Satellite System, Russian definition) etc., type.  
      Satellite radionavigation makes it possible to obtain the position of the receiver by a method much like triangulation. The distances are measured on the basis of signals sent by satellites.  
      The signals transmitted by the satellites are formed by modulating the carrier of the signal with a spreading code. Thus, the satellite signals allow two types of measurement for locating the receiver. Moreover, the modulation of the carrier by a spreading code extends the spectrum in the spectral band, thereby boosting the resistance of the system to jamming. Also, furthermore, this makes it possible to distinguish between the satellites (by using a different code per satellite).  
      The first type of distance measurement by satellite radionavigation is a conventional measurement based on the carrier of the signal received. The measurements based on the phase of the carrier are accurate but ambiguous. Thus, the receiver is capable of evaluating the number of wavelengths between the satellite and the receiver to within modulo the wavelength.  
      The second type of distance measurement uses the code of the signal received. Unlike the measurements based on the carrier, the measurements based on the code are not ambiguous, since the receiver is capable of evaluating the integer number of code periods between the satellite and the receiver. However, the measurements based on the code are much less accurate than those based on the phase.  
      To perform these two types of measurement, the receiver acquires and tracks the signal received. To do this, it generates so-called local replicas of the code and of the carrier, which it correlates with the signal received. The code and the carrier being incoherent information, the generations of the code replica and carrier replica are controlled by two distinct loops.  
      The receiver uses these two loops to obtain accurate and unambiguous measurements. In a first phase, the so-called acquisition phase, the receiver functions in open loop so as to search for the signal received by testing several hypothesis regarding position and speed of the local code and of the local carrier. Once the code search has made it possible to resolve the ambiguity, the receiver functions in closed loop. The code loop is aided by the carrier loop speed-wise making it possible to exploit the accuracy offered by the phase measurement with no limitation by the ambiguity. Thus, the dynamic of the carrier and of the clock are eliminated and the code measurement can be further filtered (smoothing of the code by the carrier) so as to improve the accuracy thereof.  
      When the code passband used is narrow, there is a risk of tailoff or even of the loss of lock of the code loop on account of the residual dynamic. When the band used is wide so as to come within the dynamic, the measurements are noisy.  
      The measurements thus obtained are marred with so-called ionospheric errors, due to the propagation through the ionosphere, the effects of which depend on frequency. This phenomenon induces measurement errors of like value but of opposite sign between the code and the carrier. There is therefore incoherence between the carrier loop and the code loop. The dynamic of these errors, small but not negligible, necessitates a minimum passband of the code loop and hence potentially limits the accuracy.  
      Single-frequency receivers do not make it possible to evaluate the ionospheric error. This error can therefore only be corrected roughly through a bias.  
      The ionospheric error depending on frequency, the use of a dual-frequency receiver makes it possible to calculate the offset between these two frequencies, to deduce the absolute ionospheric error therefrom and to correct the speed aid sent to the code loop as shown by  FIG. 1 .  
      The two signals s 1  and s 2  correspond to two distinct frequencies. Traditionally, these two signals s 1  and s 2  are processed independently. The signals s 1  (respectively s 2 ) are correlated by the carrier  130  (respectively  230 ) then by the code  140  (respectively  240 ). The signals thus obtained from the signals s 1  (respectively s 2 ) are processed by the integration and resetting device  150  (respectively  250 ).  
      At the output of the integration and resetting device  150  are obtained four signals: the signal I A   1  formed by the aggregate samples in-phase for the carrier and phase lead for the code, the signal I R   1  formed by the aggregate samples in-phase for the carrier and phase lag for the code, the signal Q A   1  formed by the aggregate samples in quadrature of the carrier and phase lead for the code, the signal Q R   1  formed by the aggregate samples in quadrature for the carrier and phase lag for the code. Also, at the output of the integration and resetting device  250  are obtained four signals: the signal I A   2  formed by the aggregate samples in-phase for the carrier and phase lead for the code, the signal I R   2  formed by the aggregate samples in-phase for the carrier and phase lag for the code, the signal Q A   2  formed by the aggregate samples in quadrature for the carrier and phase lead for the code, the signal Q R   2  formed by the aggregate samples in quadrature for the carrier and phase lag for the code.  
      The signals I A   1 , I R   1 , Q A   1 , Q R   1  (respectively I A   2 , I R   2 , Q A   2  Q R   2 ) are processed by a phase discriminator  161  (respectively  261 ) and code discriminator  162  (respectively  262 ). The information obtained by the discriminators  161  and  162  (respectively  261  and  262 ) are used by the loop corrector  170  (respectively  270 ) to deliver the carrier speed to the carrier oscillator  110  (respectively  210 ) and the code speed to the code oscillator  120  (respectively  220 ). These oscillators  110 ,  120 ,  210  and  220  are for example numerically controlled oscillators (NCO). The carrier oscillator  110  (respectively  210 ) makes it possible to generate a carrier replica used for the correlation  130  (respectively  230 ) with the signal s 1  (respectively s 2 ). The code oscillator  120  (respectively  220 ) makes it possible to generate a code replica used for the correlation  140  (respectively  240 ) with the signal s 1  (respectively s 2 ) correlated by the carrier replica. The device  300  calculates by linear combination the measurement corrected on the basis of the two measurements marred with the ionospheric error originating from the two independently processed signals.  
      Such a solution makes the measurement less robust to the dynamic if the band used is narrow and less accurate if the band used is wide.  
      The present invention makes it possible to alleviate these drawbacks, in particular the use of a narrow band allowing good accuracy while having a system robust to dynamic. The invention proposes a process of dual-frequency reception, the relative dynamic of the signals received being small, said process comprising per frequency a code loop and a carrier loop that are incoherent and at least the following steps: 
          a change of reference of the dual-frequency base to a (mean, offset) base,     a correction on at least the offsets loop so as to obtain the offsets speed in this (mean, offset) base,     an inverse change of reference so to as calculate on the basis of the offsets speed in the (mean, offset) base the relative speed in the dual-frequency base,     a correction of the code speed for each of the two frequencies by the relative speed obtained in the dual-frequency base.        

      The invention consists of a (mean, offset) converter of a dual-frequency receiver with carrier loop and incoherent code loop, allowing the change of reference of the phases of each of the frequencies to their phase mean and their phase offset if it receives information dependent on these said phases.  
      An exemplary (mean, offset) converter according to the invention receives, for each of the two frequencies, at least one signal e 1  (respectively e 2 ) originating from at least one discriminator  161 ,  162 ,  261  or  262  associated with this frequency, each of these signals being weighted by a weighting coefficient λ 1  (respectively λ 2 ) associated with the signal, and calculates the offset Δ=λ 1  e 1 −λ 2  e 2  and the mean Σ=αλ 1  e 1 −βλ 2  e 2  of these weighted signals, α and β being coefficients whose value is determined as a function of the respective incoming signals e 1  and e 2 .  
      Another subject of the invention is an inverse (mean, offset) converter  415  of a dual-frequency receiver with carrier loop and incoherent code loop, characterized in that it allows at least the obtaining of the relative speed if it receives the phase offset speed of the two frequencies.  
      An exemplary inverse (mean, offset) converter according to the invention consists of an inverse (mean, offset) converter  415  receiving the speed of the offset v e  and the speed of the mean v m  respectively from the corrector of the offsets loop  413  and from the corrector of the mean loop  414 , and calculating for each of the two frequencies the carrier speeds and/or the relative speed (respectively the code speeds) if the signals converted by the (mean, offset) converter  412  originate from a phase discriminator  161 ,  261  (respectively from a code discriminator  162 ,  262 ).  
      In one of its variants, the invention proposes a loop corrector of a dual-frequency receiver with carrier loop and incoherent code loop, comprising: 
          at least four inputs, the first two receiving the signals from the phase discriminators  161 ,  261  of the two frequencies and the following two receiving the signals from the code discriminators  162 ,  262  of the two frequencies,     at least one weighter  411 ,  416  coupled to each input, the weighting value λ being the wavelength of the signal for the first two inputs and the shift length for the following two,     a (mean, offset) converter  412  according to the invention receiving the first two weighted inputs and delivering the phase offset and mean and/or (mean, offset) converter  412  according to the invention receiving the following two weighted inputs and delivering the code offset and the mean,     coupled to the offset output of each (mean, offset) converter  412  an offsets loop  413  and to the mean output of each (mean, offset) converter  412  a mean loop  414 ,     an inverse (mean, offset) converter  415  according to the invention coupled to each of the offsets loop  413 /mean loop  414  pairs.        

      The invention consists, furthermore, of a dual-frequency receiver comprising per frequency a code loop and a carrier loop that are incoherent, said dual-frequency receiver receiving signals whose relative dynamic is small, comprising at least: 
          a (mean, offset) converter  412  according to the invention allowing the change of reference of the phases to their phase mean and their phase offset,     a phase offsets loop corrector  413  making it possible to obtain on the basis of the phase offsets emanating from the converter a phase offset speed,     an inverse (mean, offset) converter  415  according to the invention allowing the change of reference of the phase offset speed so as to obtain the relative speed,     two correctors  180  and  280  of the code speed, one per frequency, each receiving the respective code speed, carrier speed and relative speed emanating from the inverse converter, and each delivering its respective corrected code speed to its respective code loop.        

      In a first variant of the dual-frequency receiver according to the invention, the (mean, offset) converter  412  receives the carrier measurements calculated on the basis of the two frequencies.  
      In a second variant of the invention, a dual-frequency receiver comprising per frequency a code loop and a carrier loop that are incoherent, said dual-frequency receiver receiving signals whose relative dynamic is small, comprises at least: 
          a loop corrector for example as described previously delivering the relative speed and for each of the two frequencies the code speed and the carrier speed,     two code speed correctors  180  and  280 , one per frequency: 
            each receiving said code speed, carrier speed and relative speed weighted by −2/λ 2 , where λ is the wavelength associated with the frequency of the code speed corrector, and     each delivering its respective corrected code speed to respective code oscillator.   
               

    
    
      The characteristics and advantages of the invention will become more clearly apparent on reading the description, given by way of example, and of the figures pertaining thereto which represent:  
       FIG. 1 , a dual-frequency receiver for the measurement of distance with correction of the ionospheric error according to the state of the art,  
       FIG. 2 , a diagrammatic representation of an exemplary dual-frequency receiver for the measurement of distance with correction of the ionospheric error according to the invention,  
       FIG. 3 , a first variant of the dual-frequency receiver for the measurement of distance with correction of the ionospheric error according to the invention,  
       FIG. 4 , a second variant of the dual-frequency receiver for the measurement of distance with correction of the ionospheric error according to the invention,  
       FIG. 5 , a first exemplary loop corrector of the second variant of the dual-frequency receiver according to the invention,  
       FIG. 6 , a second exemplary loop corrector of the second variant of the dual-frequency receiver according to the invention. 
    
    
       FIG. 2  shows a generic diagrammatic representation of an exemplary dual-frequency receiver for the measurement of distance with correction of the ionospheric error according to the invention. The phase discriminators  161  and  261  of the two frequencies and the code discriminators  162  and  262  are coupled to the system  400  which calculates the carrier speed and code speed for each frequency as well as the ionospheric speed.  
      The ionospheric speed will correct the code speeds of each of the two frequencies so as to remove the errors induced by the propagation of the signals through the ionosphere. This correction is performed for each frequency with the aid of a code speed corrector  180  and  280 . This ionospheric speed is calculated by the system  400  not in the dual-frequency base but in the (mean, offset) base and is then translated into the dual-frequency base.  
      The distance measurements m emanating from the system  400  are thus more accurate owing to the separation of the dynamic. Specifically, the use of the (mean, offset) base allows the use of a narrow band for the offset, whose dynamic is small, so as to improve the accuracy, and to aggregate the energies for the mean so as to improve the accuracy and the robustness to jamming.  
      In its first variant represented by  FIG. 3 , for each frequency the receiver has at its disposal a loop corrector  411   1  and  412   1  within the system  400   1 . Thus, for each frequency, the discriminators  161 ,  162 ,  261  or  262  associated  161  and  162  (respectively  261  and  262 ) are coupled to the associated loop corrector  411   1  (respectively  412   1 ). A device  420   1  receives the code speed and carrier speed for each frequency since it is coupled to the outputs of the two loop correctors  411   1  and  412   1 . This device  420   1  delivers the ionospheric speed to the code speed correctors  180  and  280 .  
      The ionospheric speed and the carrier speeds originating from the loop corrector  411   1  and  412   1  associated with each of the two frequencies will correct the code speeds originating from the loop corrector  411   1  and  412   1  associated with each of the two frequencies. This code speed correction by the ionospheric speed and the associated carrier speed makes it possible to remove the errors induced by the propagation of the signals through the ionosphere. It is performed with the aid for each frequency with the aid of a code speed corrector  180  and  280 . This ionospheric speed is calculated by the device  420   1  not in the dual-frequency base but in the (mean, offset) base and then translated into the dual-frequency base.  
      For each frequency, the code speed is, furthermore, corrected by the carrier speed calculated by the respective loop corrector  411   1  and  412   1  with the aid of a code speed corrector  180  and  280 .  
      The distance measurements m emanating from the device  420   1  are therefore more accurate on account of the separation of the dynamic.  
      In its second variant represented by  FIG. 4 , the receiver has at its disposal a loop corrector  410   2  common to the two frequencies within the system  400   2 .  
      Thus, the phase discriminators  161  and  261 , and the code discriminators  162  and  262  of the two frequencies are coupled to the common loop corrector  410   2 . The phase discriminators  161  and  261  are coupled to the first two inputs of the loop corrector  410   2  and the code discriminators  162  and  262  are coupled to the following two inputs of the loop corrector  410   2 .  
      The loop corrector  410   2  calculates, on the basis of the signals thus received, the code speed and carrier speed for each frequency as well as the ionospheric speed. The ionospheric speed is weighted for each frequency by a gain  421   2  (respectively  422   2 ) within the system  400   2 , then delivered to the code speed corrector  180  (respectively  280 ). The gains  421   2  and  422   2  are equal to −2.λ 2 , where λ is the wavelength associated with each frequency.  
      The ionospheric speed at the output of the system  400   2  will correct the code speeds originating from the common loop corrector  410   2  for each of the two frequencies. This code speed correction by the ionospheric speed makes it possible to remove the errors induced by the propagation of signals through the ionosphere. It is performed with the aid for each frequency with the aid of a code speed corrector  180  and  280 . This ionospheric speed is calculated by the loop corrector  410   2  not in the dual-frequency base but in the (mean, offset) base and then translated into the dual-frequency base. For each frequency, the code speed is, furthermore, corrected by the carrier speed calculated by the loop corrector  410   2  with the aid of a code speed corrector  180  and  280 .  
      The distance measurements m emanating from the system  400   2  are thus more accurate on account of the separation of the dynamic.  
      In the first exemplary loop corrector  410   2  common to the two frequencies, proposed by  FIG. 5 , only the signals received on its first two inputs and originating from the phase discriminators  161  and  261  are transposed from the dual-frequency base to the (mean, offset) base. The four signals e 1   p , e 2   p , e 1   c , and e 2   c  received by the loop corrector  410   2  are weighted by associated weighting coefficients  411   1 ,  411   2 ,  416   1  and  416   2 .  
      These weighting coefficients  411   1 ,  411   2 ,  416   1  and  416   2  are, either the wavelength of the signal originating from the phase discriminator, or the shift length (otherwise known as “chip” length) of the signal originating from the code discriminator. Hence, the weighting coefficient  411   1  of the signal e 1   p  originating from the phase discriminator  161  associated with the first frequency has as value the wavelength λ 1   p  of this signal. The weighting coefficient  411   2  of the signal e 2   p  originating from the phase discriminator  261  associated with the second frequency has as value the wavelength λ 2   p  of this signal. Hence, the weighting coefficient  416   1  of the signal e 1   c  originating from the phase discriminator  162  associated with the first frequency has as value the wavelength λ 1   c  of this signal. The weighting coefficient  416   2  of the signal e 2   c  originating from the phase discriminator  262  associated with the second frequency has as value the wavelength λ 2   c  of this signal.  
      The signals e 1   p  and e 2   p  originating from the phase discriminators  161  and  261  are transposed to the (mean, offset) base with the aid of a (mean, offset) converter  412  receiving these weighted signals, that is to say λ 1   p  e 1   p  and λ 2   p  e 2   p . The devices  412   1  and  412   2  of the converter calculate, on the basis of the two incoming signals respectively the offset and the mean. The device  412   1  is therefore an offset calculator and the device  412   2  a mean calculator.  
      Generally, regardless of the type of discriminator  161 ,  162 ,  261  or  262  from which the signals λ 1  e 1  and λ 2  e 2  received by the (mean, offset) converter  412  originate, the output of the offset calculator  412   1  is equal to Δ=λ 1  e 1 −λ 2  e 2  and the output of the mean calculator  412   2  is equal to Σ=α λ 1  e 1 +β λ 2  e 2 , α and β being coefficients whose value is determined as a function of the respective incoming signals e 1  and e 2 .  
      In the example of  FIG. 5 , the incoming signals λ 1   p  e 1   p  and λ 2   p  e 2   p , originating from phase discriminators  161 ,  261 , the values at the output of the (mean, offset) converter  412  are the phase offset and the phase mean of two frequencies. Also, the coefficients α and β are calculated as a function of the signal-to-noise ratios estimated on the two frequencies and of the wavelength.  
      The output of the offset calculator  412   1  is coupled to an offset loop corrector  413  delivering the offset speed v e  and the output of the mean calculator  421   2  is coupled to a mean loop corrector  414  delivering the mean speed v m . These two speeds, offsets speed v e  and means speed v m , are transmitted to an (mean, offset) inverse converter  415 . The (mean, offset) inverse converter  415  transposes the speeds that it receives into the dual-frequency base.  
      In the case of  FIG. 5 , the speeds received by the (mean, offset) inverse converter  415  being the phase offsets speed v e   p  and phase means speed v m   p , the speeds at the output of the converter are the carrier speeds for the two frequencies and the ionospheric speed. The device  415   1 , dubbed subconverter f 1 , calculates the carrier speed associated with the first frequency on the basis of the phase offsets speeds v e   p  and the phase means speeds v m   p . The device  415   2 , dubbed subconverter f 2 , calculates the carrier speed associated with the second frequency on the basis of the phase offsets speeds v e   p  and phase means speeds v m   p . Also, the device  415   + , dubbed iono subconverter, calculates the ionospheric speed on the basis of the phase offsets speed v e   p .  
      In general, regardless of the type of offsets speed v e  and means speed v m  at the input of the (mean, offset) inverse converter  415 , the output from subconverter f 1  associated with α is equal to  
             1     α   +   β       ⁢     v   m       +       β     α   +   β       ⁢     v   e         ,       
 
 and the output from subconverter f 2  associated with β is equal to  
           1     α   +   β       ⁢     v   m       +       α     α   +   β       ⁢       v   e     .           
 
      In the particular case depicted by  FIG. 5 , the (mean, offset) inverse converter  415  has at its disposal an extra output on which it delivers the ionospheric speed calculated by the iono subconverter  415   +  and equal to  
         1       λ   1   2     -     λ   2   2         .       
 
      The inputs e 1   c , and e 2   c  of the loop corrector  410   2  of the example of  FIG. 5  not being transposed into the (mean, offset) base, the output of each weighter  416   1  and  416   2  is coupled to a code loop corrector  417  calculating the code speed specific to each frequency.  
      In this example, the order and the loop band for the mean loop ( 414 ) are compatible with the dynamic of the carrier (high) and of the clock. The order and the loop band for the offsets loop are compatible with the dynamic of the ionospheric error (small). By eliminating the dynamic of the ionospheric error by virtue of the code speed correctors  180  and  280  (in addition to the dynamic of the carrier by virtue of the corrections by the carrier speeds) it is possible to considerably reduce the band of the code loops and hence to gain accuracy.  
      In the second exemplary loop corrector  410   2  common to the two frequencies, proposed by  FIG. 6 , the signals received on the four inputs, originating from the phase discriminators  161  and  261  for the first two inputs and from the code discriminators  162  and  262  for the following two inputs, are transposed from the dual-frequency base into the (mean, offset) base. The four signals e 1   p , e 2   p , e 1   c , and e 2   c  received by the loop corrector  410   2  are weighted by associated weighting coefficients  411   1 ,  411   2 ,  416   1  and  416   2 .  
      These weighting coefficients  411   1 ,  411   2 ,  416   1  and  416   2  are, either the wavelength of the signal originating from the phase discriminator, or the shift length (otherwise known as “chip” length) of the signal originating from the code discriminator. Hence, the weighting coefficient  411   1  of the signal e 1   p  originating from the phase discriminator  161  associated with the first frequency has as value the wavelength λ 1   p  of this signal. The weighting coefficient  411   2  of the signal e 2   p  originating from the phase discriminator  261  associated with the second frequency has as value the wavelength λ 2   p  of this signal. Hence, the weighting coefficient  416   1  of the signal e 1   c  originating from the phase discriminator  162  associated with the first frequency has as value the shift length λ 1   c  of this signal. The weighting coefficient  416   2  of the signal e 2   c  originating from the phase discriminator  262  associated with the second frequency has as value the shift length λ 2   c  of this signal.  
      The four signals e 1   p , e 2   p , e 1   c , and e 2   c  originating respectively from the phase discriminators  161  and  261 , and from the code discriminators  162  and  262  are transposed into the (mean, offset) base with the aid of two (mean, offset) converters  412   p  et  412   c  receiving these weighted signals, that is to say λ 1   p  e 1   p , λ 1   p  e 2   p , λ 2   c  e 1   c  and λ 2   c  e 2   c . The first (mean, offset) converter  412   p  receives the signals e 1   p , e 2   p  originating respectively from the phase discriminators  161  and  261 , and the second (mean, offset) converter  412   c  the signals e 1   c , and e 2   c  originating respectively from the code discriminators  162  and  262 .  
      In general, the devices  412   1  and  412   2  of the converter calculate, on the basis of the two incoming signals respectively the offset and the mean. The device  412   1  is therefore an offset calculator and the device  412   2  a mean calculator. Regardless of the type of discriminator  161 ,  162 ,  261  or  262  from which the signals λ 1  e 1  and λ 2  e 2  received by the (mean, offset) converter  412  originate, the output of the offset calculator  412   1  is equal to Δ=λ 1  e 1 −λ 2  e 2  and the output of the mean calculator  412   2  is equal to Σ=α λ 1  e 1 +β λ 2  e 2 , α and β being coefficients whose value is determined as a function of the respective incoming signals e 1  and e 2 .  
      In the example of  FIG. 6 , for the incoming signals λ 1   p  e 1   p  and λ 2   p  e 2   p , originating from phase discriminators  161  and  261 , the values at the output of the (mean, offset) converter  412   p  are the phase offset and the phase mean of the two frequencies. Also, the coefficients α p  and β p  are calculated as a function of the signal-to-noise ratios estimated on the two frequencies and of the wavelength. Whereas for the incoming signals λ 1   c  e 1   c  and λ 2   c  e 2   c , originating from code discriminators  162  and  262 , the values at the output of the (mean, offset) converter  412   c  are the code offset and the code mean of two frequencies. Also, the coefficients α c  and β c  are calculated as function of the signal-to-noise ratios estimated on the two frequencies and of the shift length (otherwise known as “chip” lengths).  
      In general, the output of the offset calculator  412   1  is coupled to an offset loop corrector  413  delivering the offset speed v e  and the output of the mean calculator  412   2  is coupled to a mean loop corrector  414  delivering mean speed v m . These two speeds, offsets speed v e  and means speed v m , are transmitted to an (mean, offset) inverse converter  415 . The (mean, offset) inverse converter  415  transposes the speeds that it receives into the dual-frequency base.  
      In the case of  FIG. 6 , the speeds received by the (mean, offset) inverse converter  415   p  being the phase offsets speed v e   p  of the phase offsets loop  413   p  and the phase means speed v m   p  of the phase mean loop  414   p , the speeds at the output of the (mean, offset) inverse converter  415   p  are the carrier speeds for the two frequencies and the ionospheric speed; and the speeds received by the (mean, offset) inverse converter  415   c  being the code offsets speed v e   c  of the phase offsets loop  413   c  and code means speed v m   c  of the phase mean loop  414   c , the speeds at the output of the (mean, offset) inverse converter  415   c  are the code speeds for the two frequencies.  
      In general, within an (mean, offset) inverse converter  415 , the device  415   1 , dubbed subconverter f 1 , calculates the carrier speed associated with the first frequency on the basis of the offsets speeds v e  and the means speeds v m . The device  415   2 , dubbed subconverter f 2 , calculates the speed associated with the second frequency on the basis of the offsets speeds v e  and means speeds v m . Regardless of the type of offsets speed v e  and means speed v m  at the input of the (mean, offset) inverse converter  415 , the output from subconverter f 1  associated with a is equal to  
             1     α   +   β       ⁢     v   m       +       β     α   +   β       ⁢     v   e         ,       
 
 and the output from subconverter f 2  associated with β is equal to  
           1     α   +   β       ⁢     v   m       +       α     α   +   β       ⁢       v   e     .           
 
      In particular, if the speeds received by the (mean, offset) inverse converter  415   p  are phase offsets speeds v e   p  and phase means speeds v m   p , the speeds at the output are carrier speeds; if the speeds received by the (mean, offset) inverse converter  415   c  are code offset speeds v e   c  and code means speeds v m   c , the speeds at the output are code speeds.  
      Moreover, the device  415   + , dubbed iono subconverter, of the (mean, offset) inverse converter  415   p , receiving phase offsets speeds v e   p  and phase means speeds v m   p , calculates the ionospheric speed on the basis of the phase offsets speed v e   p . In the particular case depicted by  FIG. 6 , the (mean, offset) inverse converter  415  has at its disposal an extra output on which it delivers the ionospheric speed calculated by the iono subconverter  415   +  and equal to  
         1       λ   1   2     -     λ   2   2         .       
 
      Another example (not illustrated) would consist in transposing only the outputs of the code discriminators  162 ,  262  into the (mean, offset) base.  
      The ionospheric speed calculated to correct the errors related to the propagation of the signals through the ionosphere. Now, the signals of different frequencies propagating through the ionosphere have a relatively small dynamic. Hence, more generally the various systems and devices described may be applied in any dual-frequency receiver having incoherent code and carrier loop whose different frequency signals have a relatively small dynamic. The speed calculated to correct the errors inducing this small relative dynamic is dubbed the relative speed.  
      This type of dual-frequency reception system for the measurement of distance with incoherent carrier and code loop using the invention can be applied not only to GPS, to Glonass and to Galileo but also to any application requiring the use of a dual-frequency receiver with incoherent carrier and code loop and receiving signals with small relative dynamic.