Patent Publication Number: US-7710311-B2

Title: Short range radar small in size and low in power consumption and controlling method thereof

Description:
This application is a U.S. National Phase Application under 35 USC 371 of International Application PCT/JP2005/018661 filed Oct. 7, 2005. 
   TECHNICAL FIELD 
   The present invention relates to short range radars and a method for controlling thereof. In particular, the present invention relates to short range radars and a method for controlling thereof employing a technique for achieving, with a simple and small sized construction, short range radars used in the range of a quasi millimeter band (UWB: Ultra-wideband) from 22 GHz to 29 GHz allocated for automotive radars or radars for walk assistance of visually handicapped persons, in particular, and achieving low power consumption, from among short range radars for radiating pulse waves of narrow width (short range waves) to a space in a predetermined cycle, receiving and detecting a reflection wave from an object which exists in the space, and analyzing the object which exists in the space based on its detected output. 
   BACKGROUND ART 
   A pulse radar for investigating an object in space by using conventionally known pulse waves basically has a construction as shown in  FIG. 14 . 
   That is, in this pulse radar  10  shown in  FIG. 14 , upon the receipt of a trigger signal G outputted in a predetermined cycle Tg from a control section  16  described later, a transmitter section  11  generates a pulse wave Pt having a predetermined width and a predetermined carrier frequency synchronized with the trigger signal G and radiates the generated pulse wave to a space via a transmitter antenna  11   a.    
   This pulse wave Pt is reflected by means of an object  1   a  which exists in a space  1 , and its reflection wave Pr is received by a receiver antenna  12   a  of a receiver section  12 , and then, the received wave is detected by means of a detector circuit  13 . 
   A signal processor section  15  analyzes the object  1   a  which exists in the space  1  based on a timing with which a detected output D is outputted from the receiver section  12  while a timing with which a pulse wave is transmitted from the transmitter section  11  is defined as a reference timing, for example, or its outputted waveform. 
   The control section  16  makes a variety of controls with respect to the transmitter section  11  and the receiver section  12  based on a processing result or the like of the signal processor section  15 . 
   A basic construction of such pulse radars  10  is disclosed in patent documents 1 and 2 below. 
   Patent document 1: Jpn. Pat. Appln. KOKAI Publication No. 7-012921 
   Patent document 2: Jpn. Pat. Appln. KOKAI Publication No. 8-313619 
   From among the pulse radars having such a basic construction, the following two types of pulse radars are devised as automotive radars which have been practically available in recent years. 
   The development of pulse radars of a first type is underway for the purpose of assistance at the time of high speed running such as prevention of collision of automobiles or running control by investigating a narrow angle range with high output and in long distance using a millimeter wave band (77 GHz). 
   The development of pulse radars of a second type is underway for the purpose of assistance at the time of low speed running such as automobile dead angle assistance or assistance of putting a car in garage by investigating a wide angle range with low output and in long distance using a quasi millimeter wave (22 GHz to 29 GHz). 
   The quasi millimeter band for use in the pulse radars of this second type is generally referred to as an UWB (Ultra-wideband), and is used for medical radars, radars for walk assistance of visually handicapped persons, or a short distance communication system or the like as well as automotive radars. 
   The UWB is a wide bandwidth, and thus, in a radar system, a short pulse having a width shorter than 1 ns can be used, and it is expected that short range radars having high distance resolution can be achieved. 
   DISCLOSURE OF INVENTION 
   However, in actuality, there are a variety of problems to be solved, which will be described later, in order to achieve short range radars having high distance resolution using this UWB. 
   One of the important problems is that, although there is a need for downsizing and low power consumption in incorporation of automotive radars into a variety of vehicles or portable use of radars for walk assistance of visually handicapped persons, conventional pulse radars cannot respond to such a need sufficiently. 
   That is, from the fact that phase information can be obtained by a receiver section  12  of the conventional pulse radars, a quadrature type detector circuit is used as a detector circuit  13 . 
   This quadrature type detector circuit  13 , as shown in  FIG. 15 , branches an input signal S in phase by means of a distributor  13   a , and inputs the branched signals to two mixers  13   b  and  13   c , respectively. 
   Here, a local signal L is inputted to the two mixers  13   b  and  13   c , respectively, after divided into signals each having a 90-degree phase difference by means of a 90-degree distributor  13   d.    
   Then, the two mixers  13   b  and  13   c  mix the input signal S with the local signal L divided into the signals each having a 90-degree phase difference. 
   The local signal L is used to branch a part of the pulse waves (transmission waves) from the transmitter section  11  shown in  FIG. 14 , for example. 
   Then, two filters  13   e  and  13   f  sample baseband components I and Q from the output components from the two mixers  13   b  and  13   c.    
   A computing process for these baseband components I and Q is carried out by means of a signal processor section  15  shown in  FIG. 14  after processed via a sample hold circuit or an A/D converter and the like, although not shown, for example, thereby making it possible to grasp strength and a phase of an input signal S, i.e., a reflection wave Pr from the object  1   a  shown in  FIG. 14 . 
   Hence, such a quadrature type detector circuit  13  not only requires two mixers  13   b  and  13   c  but also requires two systems such as a circuit that follows these mixers, such as a sample hold circuit or an A/D converter, for example, and there is a problem that an equipment construction of the pulse radars becomes complicated, resulting in higher cost. 
   Further, the quadrature type detector circuit  13  requires an amplifier or the like because there is a need for supplying a local signal with sufficient power to the two mixers  13   b  and  13   c , and there is a problem that a whole equipment construction of pulse radars becomes complicated, resulting in high power consumption. 
   In addition, a 90-degree distributor  13   d  in a quasi millimeter band is proper in a circular “rat race” type because of its distribution constant type and a small loss. 
   Hence, there is a problem that this “rat race” type structured 90-degree distributor  13   d  is hardly hybridized with an IC circuit, and a circuit construction becomes large-sized. 
   In addition, a frequency of a local signal L for use in the quadrature type detector circuit  13  is a receiving frequency itself, and moreover, is at a high level, as described above. Thus, there is a need for heavy shielding so as to prevent the cable run or receiving of the leak component. Therefore, there is a problem that equipment downsizing becomes difficult. 
   On the other hand, it is possible to consider use of a peak detector circuit with a diode used in power measurement or the like instead of using the quadrature type detector circuit with its complicated construction and high power consumption as described above. 
   Hence, the peak detector circuit with a diode is low in response speed in principle, and cannot detect a receiving signal having a short pulse of 1 ns or less as described above. 
   In addition, in the case where a target serving as an object  1   a  has a high reflection factor such as a metal plate, a transmission pulse waveform is analogous to a receiving waveform reflected and returned from the target. 
   In this case, as described previously, the quadrature type detector circuit  13  used as a local signal by branching a transmission wave is employed as a detector circuit, a correlation of a detected output is obtained by means of a signal processor section  15 , thereby making it possible to detect a target with high sensitivity. 
   Hence, with respect to a target having dispersion property such as a human body, even if the quadrature type detector circuit  13  is employed as a detector circuit, a receiving pulse has a long tail, and its waveform is different from an ideal waveform. Thus, there is a problem a correlation output becomes small, and a radars sensing capability is lowered. 
   The present invention has been made in order to solve the above-described problems associated with the conventional technique. It is an object of the present invention to provide short range radars and a method for controlling the short range radars which are available in UWB, small sized, and low in power consumption. 
   In order to achieve the above-described objects, according to a first aspect of the present invention, there is provided a short range radar comprising: a transmitter section ( 21 ) which radiates a short range wave (Pt) to a space ( 1 ); a receiver section ( 30 ) having a detector circuit ( 33 ) composed of a branch circuit ( 34 ) which receives a reflection wave (Pr) of the short range wave (Pt) radiated to the space ( 1 ) by means of the transmitter section ( 21 ) and branches in phase a signal (R′) of the reflection wave (Pr) into first and second signals (V 1 , V 2 ), a linear multiplier ( 35 ) which linearly multiplies the first and second signals (V 1 , V 2 ) branched in phase by means of the branch circuit ( 34 ), and a low pass filter ( 36 ) which samples a baseband component from an output signal from the linear multiplier ( 35 ); a signal processor section ( 40 ) which carries out an analyzing process of an object ( 1   a ) which exists in the space ( 1 ) based on an output from the receiver section ( 30 ); and a control section ( 50 ) which makes a predetermined control with respect to at least one of the transmitter section ( 21 ) and the receiver section ( 30 ) based on an analysis result from the signal processor section ( 40 ). 
   In order to achieve the above-described objects, according to a second aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein the linear multiplier ( 35 ) of the detector circuit ( 33 ) is composed of a Gilbert mixer. 
   In order to achieve the above-described objects, according to a third aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein the receiver section ( 30 ) has a sample hold circuit ( 37 ) which carries out integration with respect to an output signal of the detector circuit ( 33 ) and holds and outputs a result of the integration. 
   In order to achieve the above-described objects, according to a fourth aspect of the present invention, there is further provided the short range radar according to the third aspect, wherein the control section ( 50 ) variably controls an integration start timing and an integration time of the sample hold circuit ( 37 ) based on a processing result from the signal processor section ( 40 ). 
   In order to achieve the above-described objects, according to a fifth aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein a plurality of sample hold circuits ( 37 A,  37 B,  37 C,  37 D) are provided as the sample hold circuit ( 37 ), and the plurality of sample hold circuits ( 37 A,  37 B,  37 C,  37 D) each carry out integration in different periods from each other with respect to the output signal from the detector circuit ( 33 ). 
   In order to achieve the above-described objects, according to a sixth aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein a power amplifier ( 25 ) which amplifies the short range wave (Pt) is provided at the transmitter section ( 21 ), a low noise amplifier ( 32 ) which amplifies a signal of the reflection wave (Pr) is provided at the receiver section ( 30 ), and the control section ( 50 ) controls a gain of at least one of the power amplifier ( 25 ) provided at the transmitter section ( 21 ) and the low noise amplifier ( 32 ) provided at the receiver section ( 30 ) so that a signal level (R′) of the reflection wave (Pr) inputted to the detector circuit ( 33 ) at the receiver section ( 30 ) is within a linear operation range of the linear multiplier. 
   In order to achieve the above-described objects, according to a seventh aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein the transmitter section ( 21 ) is provided with: a pulse generator ( 23 ) which generates a pulse signal (Pa) having a predetermined width; and an oscillator ( 24 ) which operates to oscillate only in a period in which the pulse signal (Pa) from the pulse generator ( 23 ) is inputted and outputs an output signal (Pb) as the short range wave (Pt), and stops the oscillating operation in a period in which the pulse signal (Pa) is not inputted. 
   In order to achieve the above-described objects, according to an eighth aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein the control section ( 50 ) stops power supply to the transmitter section ( 21 ) in a period in which the transmitter section ( 21 ) radiates the short range wave (Pt) to the space ( 1 ), and radiates a next short range wave (Pt) to the space ( 1 ). 
   In order to achieve the above-described objects, according to a ninth aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein the control section ( 50 ) stops power supply to the receiver section ( 30 ) in a period in which the transmitter section ( 21 ) radiates the short range wave (Pt) to the space ( 1 ), and then, radiates a next short range wave (Pt) to the space ( 1 ) except a period in which a reflection wave (Pr) relevant to the short range wave (Pt) radiated to the space ( 1 ) is received by means of the receiver section ( 30 ). 
   In order to achieve the above-described objects, according to a tenth aspect of the present invention, there is further provided the short range radar according to the first aspect, wherein first and second receiver sections ( 30 A,  30 B) are provided as the receiver section ( 30 ), each of which has first and second receiving antennas ( 31 A,  31 B) provided to be spaced from each other with a predetermined distance in order to receive the reflection wave (Pr), and the signal processor section ( 40 ) analyzes a direction of an object ( 1   a ) which exists in the space ( 1 ) based on output signals from the first and second receiver sections ( 30 A,  30 B). 
   In order to achieve the above-described objects, according to an eleventh aspect of the present invention, there is further provided the short range radar according to the second aspect, wherein the Gilbert mixer used as the linear multiplier ( 35 ) of the detector circuit ( 33 ) comprises: a first differential amplifier ( 35   a ) comprising first and second transistors (Q 1 , Q 2 ) each having a base input end, a collector output end, and an emitter common current path, the emitter common current path of the first and second transistors (Q 1 , Q 2 ) being connected to a constant current source (I 1 ); a second differential amplifier ( 35   b ) comprising third and fourth transistors (Q 3 , Q 4 ) each having a base input end, a collector output end, and an emitter common current path, the emitter common current path of the third and fourth transistors (Q 3 , Q 4 ) being connected to a collector output end of the first transistor (Q 1 ) of the first differential amplifier ( 35   a ); a third differential amplifier ( 35   c ) comprising fifth and sixth transistors (Q 5 , Q 6 ) each having a base input end, a collector output end, and an emitter common current path, the base input end of the fifth transistor (Q 5 ) being connected in common to the base input end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ), the emitter common current path of the fifth and sixth transistors (Q 5 , Q 6 ) being connected to a collector output end of the second transistor (Q 2 ) of the first differential amplifier ( 35   a ); a first load resistor (R 3 ) and a first output end (OUT 1 ) connected in common to a collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and a collector output end of the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ), respectively; a second load resistor (R 4 ) and a second output end (OUT 2 ) connected in common to a collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and a collector output end of the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ), respectively; a first low pass filter (LPF 1 ) including first and second coils (L 1 , L 2 ) and a first resistor (R 9 ) and a second low pass filter (LPF 2 ) including third and fourth coils (L 3 , L 4 ) and a second resistor (R 10 ) connected in series, respectively, between a first pair of lines (+, −) and an earth line which transmits the first signal (V 1 ) branched in phase by means of the branch circuit ( 34 ); a third low pass filter (LPF 3 ) including fifth and sixth coils (L 5 , L 6 ) and a third resistor (R 11 ) and a fourth low pass filter (LPF 4 ) including seventh and eighth coils (L 7 , L 8 ) and a fourth resistor (R 12 ) connected in series, respectively, between a second pair of lines (+, −) and an earth line which transmits the second signal (V 2 ) branched in phase by means of the branch circuit ( 34 ); first and second emitter follower circuits (EF 1 , EF 2 ) comprising seventh and eighth transistors (Q 7 , Q 8 ) each having a base input end and an emitter output end, the base input ends of the seventh and eighth transistors (Q 7 , Q 8 ) each being connected to each of connecting neutral points of the first and second coils (L 1 , L 2 ) and the third and fourth coils (L 3 , L 4 ) as each of the output ends of the first and second low pass filters (LPF 1 , LPF 2 ); third and fourth emitter follower circuits (EF 3 , EF 4 ) comprising ninth and tenth transistors (Q 9 , Q 10 ) each having a base input end and an emitter output end, the base input ends of the ninth and tenth transistors (Q 9 , Q 10 ) each being connected to each of connecting neutral points of the fifth and sixth coils (L 5 , L 6 ) and the seventh and eighth coils (L 7 , L 8 ) as each of the output ends of the third and fourth low pass filters (LPF 3 , LPF 4 ); a fifth low pass filter (LPF 5 ) composed of: a ninth coil (L 9 ) connected between a common collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ) and the first load resistor (R 3 ); a tenth coil (L 10 ) connected between the common collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ) and the first output end (OUT 1 ); and the first load resistor (R 3 ); and a sixth low pass filter (LPF 6 ) composed of: an eleventh coil (L 11 ) connected between a common collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) and the second load resistor (R 4 ); a twelfth coil (L 12 ) connected between a common collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) and the second output end (OUT 2 ); and the second load resistor (R 4 ), wherein each of the base input ends of the first and second transistors (Q 1 , Q 2 ) of the first differential amplifier ( 35   a ) is connected to each of the output ends of the first and second emitter follower circuits (EF 1 , EF 2 ), respectively, and thereby the first signal (V 1 ) branched in phase by means of the branch circuit ( 34 ) is inputted to the first differential amplifier ( 35   a ); and each of the base input ends of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) is connected to each of the output ends of the third and fourth emitter follower circuits (EF 3 , EF 4 ), respectively, and thereby the second signal (V 2 ) branched in phase by means of the branch circuit ( 34 ) is inputted to the second and third differential amplifiers ( 35   b ,  35   c ), and thereby a linearly multiplied outputs of the first and second signals (V 1 , V 2 ) can be led out from at least one of the first and second output ends (OUT 1 , OUT 2 ). 
   In order to achieve the above-described objects, according to a twelfth aspect of the present invention, there is provided a short range radar controlling method comprising the steps of: preparing a transmitter section ( 21 ), a receiver section ( 30 ), and a linear multiplier ( 35 ); radiating a short range wave (Pt) to a space ( 1 ) by means of the transmitter section ( 21 ); receiving a reflection wave (Pr) of the short range wave (Pt) radiated to the space ( 1 ) by means of the receiver section ( 30 ) to branch in phase a signal (R′) of the reflection wave (Pr) into first and second signals (V 1 , V 2 ); linearly multiplying the first and second signals (V 1 , V 2 ) by means of the linear multiplier ( 35 ) to output a linearly multiplied signal; sampling a baseband component from an output signal of the linear multiplier; carrying out an analyzing process of an object ( 1   a ) which exists in the space ( 1 ) based on the baseband component; and making a predetermined control with respect to at least one of the transmitter section ( 21 ) and the receiver section ( 30 ) based on a result of the analyzing process. 
   In order to achieve the above-described objects, according to a thirteenth aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein the step of outputting the linearly multiplied signal comprises the step of carrying out linear multiplication for outputting the linearly multiplied signal by using a Gilbert mixer as the linear multiplier ( 35 ). 
   In order to achieve the above-described objects, according to a fourteenth aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, further comprising the step of, before the step of carrying out the analyzing process, carrying out integration with respect to the baseband component and holding and outputting a result of the integration. 
   In order to achieve the above-described objects, according to a fifteenth aspect of the present invention, there is further provided the short range radar controlling method according to the fourteenth aspect, wherein the step of carrying out integration with respect to the baseband component comprises the step of variably controlling a start timing of integration and an integration time with respect to the baseband component based on the result of the analyzing process. 
   In order to achieve the above-described objects, according to a sixteenth aspect of the present invention, there is further provided the short range radar controlling method according to the fourteenth aspect, wherein the step of carrying out integration with respect to the baseband component comprises the step of carrying out integration in a plurality of periods different from each other with respect to the baseband component by using a plurality of sample hold circuits ( 37 ). 
   In order to achieve the above-described objects, according to a seventeenth aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein a power amplifier ( 25 ) which amplifies the short range wave (Pt) is provided at the transmitter section ( 21 ), a low noise amplifier ( 32 ) which amplifies a signal (R) of the reflection wave (Pr) is provided at the receiver section ( 30 ), and the step of making the predetermined control comprises a step of controlling a gain of at least one of the power amplifier ( 25 ) provided at the transmitter section ( 21 ) and the low noise amplifier ( 32 ) provided at the receiver section ( 30 ) so that a signal (R′) level of the reflection wave (Pr) at the receiver section ( 30 ) is within a linear operation range of the linear multiplier ( 35 ). 
   In order to achieve the above-described objects, according to an eighteenth aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein the step of radiating a short range wave (Pt) to a space ( 1 ) by means of the transmitter section ( 21 ) comprises the steps of: generating a pulse signal (Pa) having a predetermined width; making an oscillation operation only in a period in which the pulse signal (Pa) is inputted to output an output signal (Pb) as the short range wave (Pt); and stopping an oscillation operation during a period in which the pulse signal (Pa) is not inputted so as not to output an output signal (Pb) as the short range wave (Pt). 
   In order to achieve the above-described objects, according to a nineteenth aspect of the present invention, there is further provided the short pulse radar controlling method according to the twelfth aspect, wherein the step of making the predetermined control comprises the step of: stopping power supply to the transmitter section ( 21 ) in a period in which the transmitter section ( 21 ) radiates the short range wave (Pt) to the space ( 1 ), and then, radiates a next short range wave (Pt) to the space ( 1 ). 
   In order to achieve the above-described objects, according to a twentieth aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein the step of making the predetermined control comprises the step of; stopping power supply to the receiver section ( 30 ) in a period in which the transmitter section ( 21 ) radiates the short range wave (Pt) to the space ( 1 ), and then, radiates a next short range wave (Pt) to the space ( 1 ) except a period in which a reflection wave (Pr) with respect to the short range wave (Pt) radiated to the space ( 1 ) is received by means of the receiver section ( 30 ). 
   In order to achieve the above-described objects, according to a twenty-first aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein first and second receiver sections ( 30 A,  30 B) are provided as the receiver section ( 30 ), each of which has first and second receiving antennas ( 31 A,  31 B) provided to be spaced from each other with a predetermined distance in order to receive the reflection wave (Pr), and the step of carrying out the analyzing process comprises the step of analyzing a direction of an object ( 1   a ) which exists in the space ( 1 ) based on output signals from the first and second receiver sections ( 30 A,  30 B). 
   In order to achieve the above-described objects, according to a twenty-second aspect of the present invention, there is further provided the short range radar controlling method according to the twelfth aspect, wherein, in the step of outputting the linearly multiplied signal, the Gilbert mixer used as the linear multiplier ( 35 ) comprises: a first differential amplifier ( 35   a ) comprising first and second transistors (Q 1 , Q 2 ) each having a base input end, a collector output end, and an emitter common current path, the emitter common current path of the first and second transistors (Q 1 , Q 2 ) being connected to a constant current source (I 1 ); a second differential amplifier ( 35   b ) comprising third and fourth transistors (Q 3 , Q 4 ) each having a base input end, a collector output end, and an emitter common current path, the emitter common current path of the third and fourth transistors (Q 3 , Q 4 ) being connected to a collector output end of the first transistor (Q 1 ) of the first differential amplifier ( 35   a ); a third differential amplifier ( 35   c ) comprising fifth and sixth transistors (Q 5 , Q 6 ) each having a base input end, a collector output end, and an emitter common current path, the base input end of the fifth transistor (Q 5 ) being connected in common to the base input end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ), the emitter common current path of the fifth and sixth transistors (Q 5 , Q 6 ) being connected to a collector output end of the second transistor (Q 2 ) of the first differential amplifier ( 35   a ); a first load resistor (R 3 ) and a first output end (OUT 1 ) connected in common to a collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and a collector output end of the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ), respectively; a second load resistor (R 4 ) and a second output (OUT 2 ) end connected in common to a collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and a collector output end of the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ), respectively; a first low pass filter (LPF 1 ) including first and second coils (L 1 , L 2 ) and a first resistor (R 9 ) and a second low pass filter (LPF 2 ) including third and fourth coils (L 3 , L 4 ) and a second resistor (R 10 ) connected in series, respectively, between a first pair of lines (+, −) and an earth line which transmits the first signal (V 1 ) branched in phase by means of the branch circuit ( 34 ); a third low pass filter (LPF 3 ) including fifth and sixth coils (L 5 , L 6 ) and a third resistor (R 11 ) and a fourth low pass filter (LPF 4 ) including seventh and eighth coils (L 7 , L 8 ) and a fourth resistor (R 12 ) connected in series, respectively, between a second pair of lines (+, −) and an earth line which transmits the second signal (V 2 ) branched in phase by means of the branch circuit ( 34 ); first and second emitter follower circuits (EF 1 , EF 2 ) comprising seventh and eighth transistors (Q 7 , Q 8 ) each having a base input end and an emitter output end, the base input ends of the seventh and eighth transistors (Q 7 , Q 8 ) each being connected to each of connecting neutral points of the first and second coils (L 1 , L 2 ) and the third and fourth coils (L 3 , L 4 ) as each of the output ends of the first and second low pass filters (LPF 1 , LPF 2 ); third and fourth emitter follower circuits (EF 3 , EF 4 ) comprising ninth and tenth transistors (Q 9 , Q 10 ) each having a base input end and an emitter output end, the base input ends of the ninth and tenth transistors (Q 9 , Q 10 ) each being connected to each of connecting neutral points of the fifth and sixth coils (L 5 , L 6 ) and the seventh and eighth coils (L 7 , L 8 ) as each of the output ends of the third and fourth low pass filters (LPF 3 , LPF 4 ); a fifth low pass filter (LPF 5 ) composed of: a ninth coil (L 9 ) connected between a common collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ) and the first load resistor (R 3 ); a tenth coil (L 10 ) connected between the common collector output end of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the fifth transistor (Q 5 ) of the third differential amplifier ( 35   c ) and the first output end (OUT 1 ); and the first load resistor (R 3 ); and a sixth low pass filter (LPF 6 ) composed of: an eleventh coil (L 11 ) connected between a common collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) and the second load resistor (R 4 ); a twelfth coil (L 12 ) connected between the common collector output end of the fourth transistor (Q 4 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) and the second output end (OUT 2 ); and the second load resistor (R 4 ), wherein 
   each of the base input ends of the first and second transistors (Q 1 , Q 2 ) of the first differential amplifier ( 35   a ) is connected to each of the output ends of the first and second emitter follower circuits (EF 1 , EF 2 ), respectively, and thereby the first signal (V 1 ) branched in phase by means of the branch circuit ( 34 ) is inputted to the first differential amplifier ( 35   a ), 
   each of the base input ends of the third transistor (Q 3 ) of the second differential amplifier ( 35   b ) and the sixth transistor (Q 6 ) of the third differential amplifier ( 35   c ) is connected to each of the output ends of the third and fourth emitter follower circuits (EF 3 , EF 4 ), respectively, and thereby the second signal (V 2 ) branched in phase by means of the branch circuit ( 34 ) is inputted to the second and third differential amplifiers ( 35   b ,  35   c ), and thereby a linearly multiplied outputs of the first and second signals (V 1 , V 2 ) can be led out from at least one of the first and second output ends (OUT 1 , OUT 2 ). 
   With the above-described construction, according to short range radars and controlling method thereof of the present invention, a detector circuit multiplies signals obtained by branching received reflection wave signals by a branch circuit by means of a linear multiplier to obtain its square component, and samples a baseband component from its square component by means of a filter, thereby detecting a reflection wave signal. Thus, there is no need for a local signal for detection, and concurrently, its construction is simplified, making it possible to contribute to achievement of short range radars which are small in size and low in power consumption. 
   In addition, the short range radars and controlling method thereof of the present invention is a system of integrating power of received waves unlike a conventional correlating process, and thus, is suitable for detecting a target having a so-called large dispersion property in which a transmission pulse and a receiving pulse are greatly different from each other in waveform, such as a human body. 
   Further, according to the short range radars and controlling method thereof of the present invention, an oscillator for making an oscillating operation only during a period in which a pulse is inputted and outputting a short range wave as a transmission wave is used in a transmitter section, thereby preventing the generation of the residual carrier. 
   When a reflection wave signal is detected, there occurs a problem such as unstable characteristics due to a transient response when a local signal is intermittently generated, in the conventional quadrature detecting system. However, the present invention is directed to a square detecting system whose detecting characteristics does not basically depend on a transmission waveform, and can be applied smoothly without any problem when the above-described reflection wave signal is detected. 
   That is, according to short range radars and control method thereof of the present invention, as described above, a short pulse generating system and a square detecting system in which the residual carrier is not generated are combined with each other, thereby making it possible to contribute to achievement of short range radars suitable for detection of a target having a variety of scattering characteristics with a simple construction. 

   
     BRIEF DESCRIPTION OF DRAWINGS 
       FIG. 1  is a block diagram depicting a system configuration of a first embodiment of short range radars according to the present invention; 
       FIG. 2  is a block diagram depicting an example of a transmitter for use in a transmitter section of the short range radars according to the first embodiment shown in  FIG. 1 ; 
       FIG. 3  is a view showing a pulse signal Pa of a cycle Tg inputted to a transmitter and a signal Pb formed in a rectangular shape outputted in a burst shape from the transmitter for the purpose of a description of operation of the transmitter shown in  FIG. 2 ; 
       FIG. 4  is a block diagram depicting another example of a transmitter for use in a transmitter section of the short range radars according to the first embodiment shown in  FIG. 1 ; 
       FIG. 5A  is a circuit schematic view depicting a basic type of a Gilbert mixer employed as an example of a linear multiplier of a detector circuit for use in a receiver section of the short range radars according to the first embodiment shown in  FIG. 1 ; 
       FIG. 5B  is a circuit schematic view depicting an improved type of the Gilbert mixer shown in  FIG. 5A ; 
       FIG. 6  is a view showing a sine wave shaped signal S (t) inputted in a burst shape in phase to the Gilbert mixer and a square wave S (t) 2  and its envelope (baseband) W outputted from the Gilbert mixer for the purpose of a description of operation of the Gilbert mixer shown in  FIGS. 5A and 5B ; 
       FIG. 7  is a view showing a measurement result of frequency characteristics of a linear multiplier in the case where the Gilbert mixer shown in  FIG. 5B  is employed; 
       FIG. 8  is a view showing an observed waveform of a baseband component W obtained when an output of the linear multiplier in response to an input signal of a pulse width 1 ns in the case where the Gilbert mixer shown in  FIG. 5B  is employed is subjected to a 7 GHz bandwidth limitation by means of a low pass filter; 
       FIG. 9  is a view showing a measurement result of input and output characteristics of the linear multiplier in the case where the Gilbert mixer shown in  FIG. 5B  is employed; 
       FIG. 10  is a view showing a principal construction of a sample hold circuit for use in a receiver section of the short range radars according to the first embodiment shown in  FIG. 1 ; 
       FIG. 11  is a timing chart adopted to explain an operation of the short range radars according to the first embodiment shown in  FIG. 1 ; 
       FIG. 12  is a block diagram depicting a construction of essential portions of a second embodiment of short range radars according to the present invention; 
       FIG. 13  is a block diagram depicting a construction of essential portions of a third embodiment of short range radars according to the present invention; 
       FIG. 14  is a block diagram depicting a basic construction of conventional pulse radars; 
       FIG. 15  is a block diagram depicting a basic construction of a quadrature type detector circuit for use in the conventional pulse radars shown in  FIG. 14 ; and 
       FIG. 16  is a view showing a spectrum mask of a quasi millimeter wave band UWB and a desired use frequency band (recommended bandwidth). 
   

   BEST MODE FOR CARRYING OUT THE INVENTION 
   Hereinafter, some embodiments of short range radars according to the present invention will be described with reference to the accompanying drawings. 
   First Embodiment 
   First a description will be given with respect to a construction of short range radars according to a first embodiment of the present invention. 
     FIG. 1  is a block diagram depicting a construction of a short range radar  20  according to the first embodiment of the present invention. 
   The short range radar  20  according to the present invention basically includes: a transmitter section  21  which radiates a short pulse Pt to a space  1 ; a receiver section  30  having a detector circuit  33  composed of a branch circuit  34  which receives a reflection wave Pr of a short range wave Pt radiated to the space  1  by means of this transmitter section  21  and branches in phase a signal R′ of the reflection wave Pr into first and second signals V 1  and V 2 , a linear multiplier  35  which linearly multiplies the first and second signals V 1  and V 2  branched in phase by means of this branch circuit  34 , and a low pass filter  36  which samples a baseband component from an output signal from this linear multiplier  35 ; a signal processor section  40  which carries out an analyzing process of an object  1   a  which exists in the space  1  based on an output from this receiver section  30 ; and a control section  50  which makes predetermined control with respect to at least one of the transmitter section  21  and the receiver section  30  based on an analysis result from this signal processor section  40 . 
   In addition, a method for controlling short range radars according to the present invention basically includes the steps of: preparing the transmitter section  21 , the receiver section  30  and the linear multiplier  35 ; radiating the short range wave Pt to the space  1  by the transmitter section  21 ; receiving the reflection wave Pr of the short range wave Pt radiated to the space  1  by means of this receiver section  30  and branching in phase the signal R′ of the reflection wave Pr into the first and second signals V 1  and V 2 ; linearly multiplying the first and second signals V 1  and V 2  by the linear multiplier  35  to output a linear multiplying signal; sampling a baseband component from this linear multiplied output signal; carrying out an analyzing process of the object  1   a  which exists in the space  1  based on this baseband component; and making predetermined control with respect to at least one of the transmitter section  21  and the receiver section  30  based on this analysis result. 
   Specifically, this short range radar  20  shown in  FIG. 1  is composed of the transmitter section  21 ; the receiver section  30 ; an analog/digital (A/D) converter  30 ; the signal processor section  40 ; and the control section  50 . 
   Every time the transmitter section  21  receives a trigger signal G outputted from the control section  50  at a predetermined cycle Tg, this transmitter section radiates to the space  1  via a transmitter antenna  22  a short range wave Pt having a predetermined carrier frequency Fc (for example, 26 GHz) at a predetermined bandwidth Tp (for example, 1 ns) generated as described later. 
   The transmitter antenna  22  may be shared with a receiver antenna  31  of the receiver section  30  described later. 
   This transmitter section  21 , as shown in  FIG. 1 , has: a pulse generator  23  which generates a pulse signal Pa having a bandwidth Tp synchronized with a trigger signal G from the control section  50 ; an oscillator  24  which oscillates and outputs a signal having a predetermined carrier frequency Fc in a duration Tp in which a pulse signal Pa is received from this pulse generator  23 ; a power amplifier  25  which amplifies an output signal from this oscillator  24 ; a band rejection filter (BRF)  26  which suppresses a out-of-bandwidth unnecessary radiation in response to an output signal from this power amplifier  25 ; and the transmitter antenna  22  to which a signal having passed through this BRF  26  is supplied as a transmission wave. 
   Here, some configurations of the oscillator  24  are considered. 
     FIG. 2  is a block diagram depicting an example of a configuration of the oscillator  24  for use in the transmitter section  21  of the short range radars according to the first embodiment shown in  FIG. 1 . 
   That is, this oscillator  24 , as shown in  FIG. 2 , has: a 2-input, 2-output type gate circuit  24   a  in which common input AND and NAND circuits are integrated with each other; first and second input buffers  24   b  and  24   c  of emitter follower type connected to an input section of this gate circuit  24   a ; and a delay circuit  24   e  which delays by a predetermined delay time inverted outputs of an output buffer  24   d  connected to an output section of the gate circuit  24   a  and the gate circuit  24   a  and inputs the delayed inverted outputs to the first input buffer  24   b.    
   This delay circuit  24   e  is composed of a strip line or the like, for example. 
   From the thus configured oscillator  24 , as shown in  FIG. 3A , while a pulse signal Pa having a cycle Tg is inputted to the input buffer  24   c , as shown in  FIG. 3B , a rectangular wave signal Pb having a predetermined frequency (carrier frequency) is oscillated and outputted in a burst shape. 
   A frequency of an output signal Pb from this oscillator  24  is determined depending on a total of a delay time between an input and an output of the input buffer  24   b  and the gate circuit  24   a  and a delay time of the delay circuit  24   e.    
   Here, the delay time between an input and an output of the input buffer  24   b  and the gate circuit  24   a  is a fixed value generally determined depending on a circuit device. 
   Therefore, a construction is provided so as to vary some of the constants of the delay circuit  24   e , and these constants are adjusted, and thereby an oscillation frequency of the output signal Pb of the oscillator  24  is set at a substantially center frequency (for example, 26 GHz) of the UWB. 
     FIG. 4  is a block diagram depicting another example of a configuration of the oscillator  24  for use in the transmitter section  21  of the short range radars according to the first embodiment shown in  FIG. 1 . 
   That is, the oscillator  24  according to this example of configuration, as shown in  FIG. 4 , has an amplifier  24   f ; a resonator  24   g  serving as a load of this amplifier  24   f ; and a feedback circuit  24   h  which positively feeds back an output of the amplifier  24   f  to an input side to form an oscillator circuit which operates to oscillate at a resonance frequency (for example, 26 GHz) of the resonator  24   g.    
   Further, in this oscillator  24  according to this example of configuration, a switch  24   i , the switching operation of which can be controlled by means of a pulse signal Pa, is provided between an input side (or output side) of the amplifier  24   f  and an earth line. 
   This oscillator  24  according to this example of configuration operates to oscillate when the switch  24   i  is opened in a duration in which a pulse signal Pa is inputted. In addition, in a duration in which no pulse signal Pa is inputted, the switch  24   i  is closed, and one end of a feedback loop is short-circuited in an earth line, and thereby an oscillation operation stops. 
   Here, a configuration is provided such that short-circuit and opening are established between an input side of the amplifier  24   f  and the earth line by means of the switch  24   i.    
   Hence, a configuration may be provided such that short-circuit and opening are established between an output side of the amplifier  24   f  and the earth line by means of the switch  24   i.    
   The transmitter section  21  using the oscillator  24  according to any of these configurations shown in  FIGS. 2 and 4  is configured to control an oscillating operation itself of the oscillator  24  by means of a pulse signal Pa. Thus, no carrier leakage occurs in principle. 
   Therefore, when an UWB is used, limitation of power density regulated as described later may be considered only with respect to momentary power of a short range wave outputted at the time of oscillation. Thus, transmission wave power can be efficiently used to the maximum within a limit of power density regulated in accordance with a UWB standard concurrently because no carrier leakage occurs. 
   The above-described configurations of the oscillator  24  shown in  FIGS. 2 and 4  each are provided as an example. With another circuit configuration, for example, by turning on and off power (current source or the like) of an oscillator circuit in response to a pulse signal Pa as well, a burst wave free of carrier leakage as described above can be obtained. 
   In order to obtain this burst wave, conventionally, there is used an amplification shift keying (ASK) system for pulse-modulating (ON/OFF) a 24 GHz carrier signal (continuous wave) by using a switch. 
   Hence, in such a conventional ASK system, isolation at the time of switching OFF is not complete, and a carrier leakage occurs. Moreover, in short range radars, an OFF time is overwhelmingly longer by several thousand times to several ten thousand times than ON time (for example 1 ns). Thus, even if a slight carrier leakage occurs, the large residual carrier power is generated as a whole. 
   This residual carrier limits substantial receiving sensitivity of a reflection wave with respect to a transmission wave of short range radars, thus narrowing a radar investigation range and making it difficult to detect an obstacle having a low reflection factor. 
   In addition, with respect to the UWB radar system, FCC (Federal Communication Committee) regulates in the following non-patent document 1 that average power density in a bandwidth of 22 GHz to 29 GHz be −41 dBm/MHz or less, and peak power density be 0 dBm/50 MHz or less. 
   Non-patent document 1 FCC02-08, New Part 15 Rules, “FIRST REPORT AND ORDER” 
   Namely, in the above-described UWB radar system, a total amount of energy in the bandwidth of 22 GHz to 29 GHz is regulated. Thus, if the residual carrier is large, an output level of a transmission wave must be set to be low concurrently, and an investigation distance or the like is greatly limited. 
   In order to solve this problem, as indicated by dashed line from an UWB recommended bandwidth indicated by solid line in  FIG. 16 , a center frequency of the transmission wave of short range radars is saved to a band having a narrow bandwidth (Short Range Device: SRD) from 24.05 GHz to 24.25 GHz allocated for Doppler radars, and thereby it is considered that regulations on the residual carrier by FCC can be avoided. 
   However, in this case, as shown in  FIG. 16 , there is a radiation restricted band by RR (International Radio Communication Rules) for protecting a passive sensor of EESS (Earth Survey Satellite) near SRD, and serious interference with this radiation restricted band is an interest of concern. 
   In contrast, in the present invention, as described above, a system of controlling ON/OFF an oscillating operation itself by means of a pulse signal Pa to principally prevent the generation of the residual carrier is employed as a configuration of the oscillator  24 , and thereby a frequency of a radar transmission wave can be freely set within a recommended bandwidth of a spectrum mask regulated as shown in  FIG. 16 . 
   Moreover, in the present invention, the frequency of the transmission wave can be set so as to sufficiently avoid interference with the radiation restricted band as described above. 
   A signal Pb outputted from the oscillator  24  as described above is amplified by means of the power amplifier  25 , and the amplified signal is supplied to the transmitter antenna  22  as a short range wave Pt having a predetermined carrier frequency Fc (for example, 26 GHz) via the BRF  26 . 
   In this manner, from the transmitter antenna  22 , the short range wave Pt is radiated to the space  1  targeted for investigation. 
   A gain of the power amplifier  25  can be variably controlled by means of the control section  50 . 
   On the other hand, the receiver section  30  receives a reflection wave Pr from the object  1   a  of the space  1  via the receiving antenna  31 ; amplifies a signal R of the reflection Pr by means of an LNA (Low Noise Amplifier)  32 ; and then, detects by means of a detector circuit  33  a signal R′ of the reflection wave Pr bandwidth-limited by means of a band pass filter (BPF)  41  having a bandwidth of about 2 GHz. 
   A gain of the LNA 32  can be variably controlled by means of the control section  50 . 
   The detector circuit  33  is composed of: the branch circuit  34  which branches the signal R′ of the reflection wave Pr outputted from the BRF  41  into the first signal V 1  and the second signal V 2  in phase (0 degree); the linear multiplier  35  which linearly multiplies the signals branched into two signals in that phase, i.e., the first signal V 1  and the second signal V 2 ; and a low pass filter (LPF)  36  which samples a baseband component W from an output signal of this linear multiplier  36 . 
   The linear multiplier  35  includes some systems such as use of a double balancing mixer, and a method for configuring the multiplier by using a Gilbert mixer is considered as that which operates at a high speed. 
   This Gilbert mixer, as shown in  FIG. 5A , basically consists of first to third differential amplifiers  35   a ,  35   b , and  35   c.    
   Then, the first signal V 1  is differentially inputted to a first differential amplifier  35   a  and the second signal V 2  is differentially inputted to second and third differential amplifiers  35   b  and  35   c  connected to a load side of this first differential amplifier  35   a . In this manner, only a linearly multiplied signal component—(V 1 ×V 2 ) of an inverted phase equal to a product of the first signal V 1  and the second signal V 2  and a linearly multiplied signal component (V 1 ×V 2 ) of a positive phase are outputted from common load resistors R 3  and R 4  of the second and third differential amplifiers  35   b  and  35   c.    
   Specifically, in this Gilbert mixer, the first differential amplifier  35   a  includes first and second transistors Q 1  and Q 2  each having a base input end, a collector output end, and an emitter common current path, wherein each of the base input ends of the first and second transistors Q 1  and Q 2  is connected to a first signal source V 1  and the emitter common current path is connected to an earth line in series via a constant current source I 1  and a first bias power source Vb 1 . 
   The emitter common current path of the first and second transistors Q 1  and Q 2  is lead out from a connection neutral point of emitter resistors R 1  and R 2  and a base input end of the second transistor Q 2  is connected to an earth line via a second bias power source Vb 2 . 
   In addition, the second differential amplifier  35   b  includes third and fourth transistors Q 3  and Q 4  each having a base input end, a collector output end, and an emitter common current path, wherein each of the base input ends of the third and fourth transistors Q 3  and Q 4  is connected to a second signal source V 2  and the emitter common current path of the third and fourth transistors Q 3  and Q 4  is connected to the collector output end of the first transistor Q 1  of the first differential amplifier  35   a.    
   In addition, the third differential amplifier  35   a  includes fifth and sixth transistors Q 5  and Q 6  each having a base input end, a collector output end, and an emitter common current path, wherein each of the base input ends of the fifth and sixth transistors Q 5  and Q 6  is connected to the second signal source V 2  and the emitter common current path of the fifth and sixth transistors Q 5  and Q 6  is connected to the collector output end of the second transistor Q 2  of the first differential amplifier  35   a.    
   The base input end of each of the fourth transistor Q 4  of the second differential amplifier  35   b  and the fifth transistor Q 5  of the third differential amplifier  35   c  is connected in common, and is connected to an earth line via a third bias power source Vb 3 . 
   In addition, a collector output end of the third transistor Q 3  of the second differential amplifier  35   b  and a collector output end of the fifth transistor Q 5  of the third differential amplifier  35   c  are connected to an earth line via a load resistor R 3  in common, and is connected to a first output end OUT 1 . 
   In addition, a collector output end of the fourth transistor Q 4  of the second differential amplifier  35   b  and a collector output end of the sixth transistor Q 6  of the third differential amplifier  35   c  are connected to an earth line via a load resistor R 4  in common, and is connected to a second output end OUT 2 . 
   In this manner, at least one of the linearly multiplied outputs—(V 1 ×V 2 ) and (V 1 ×V 2 ) of the first and second signals V 1  and V 2  can be lead out from the first and second output ends OUT 1  and OUT 2 . 
   As the first and second signals V 1  and V 2 , when a sine wave shaped signal S (t) as shown in  FIG. 6A , for example, is inputted to the thus configured linear multiplier  35  using the Gilbert mixer in a burst shape in phase, the output signal is produced as a waveform (S (t) 2 ) obtained by squaring the input signal S (t), as shown in  FIG. 6B , and the envelope (baseband) W is proportional to power of the input signal S (t). 
   In this way, the linear multiplier  35  using the Gilbert mixer which consists of a plurality of differential amplifiers for use in the detector circuit  33  can be configured to be very small-sized with a microwave monolithic integrated circuit (MMC). Moreover, there is no need for supplying a local signal unlike a conventional quadrature type detector circuit, and thus, power consumption is reduced concurrently. 
   In the meantime, the response characteristics of the linear multiplier  35  using the Gilbert mixer which has a basic circuit configuration as shown in  FIG. 5A  have a room to be improved for use in UWB. 
   Therefore, the inventors have improved its response characteristics by making improvement so as to carry out impedance matching or peaking correction and the like of an input/output section of a linear multiplier using the Gilbert mixer which has a basic circuit configuration as shown in  FIG. 5A , and has achieved a linear multiplier which can be fully used in UWB. 
     FIG. 5B  shows a circuit configuration of a Gilbert mixer of improved type achieved by the inventors. 
   In  FIG. 5B , like constituent elements of the Gilbert mixer having a basic circuit configuration shown in  FIG. 5A  are designated by like reference numerals. A duplicate description is omitted here. 
   That is, as shown in  FIG. 5B , in the Gilbert mixer of improved type, an emitter common current path of the third and fourth transistors Q 3  and Q 4  of the second differential amplifier  35   b  is lead out from a connection neutral point of the emitter resistors R 5  and R 6 . In addition, the emitter common current path of the fifth and sixth transistors Q 5  and Q 6  of the third differential amplifier  35   c  is lead out from a connection neutral point of the emitter resistors R 7  and R 8 . 
   Although use of these pairs of emitter resistors R 5  and R 6 , and R 7  and R 8  is desirable in principle, as in the emitter resistors R 1  and R 2  of the first and second transistors Q 1  and Q 2  of the first to third differential amplifiers  35   a , not so serious problem occurs in an actual circuit configuration even if they are eliminated. 
   In addition, in the Gilbert mixer of improved type as shown in  FIG. 5B , first to fourth low pass filters LPF 1 , LPF 2 , LPF 3 , and LPF 4  and first to forth emitter follower circuits EF 1 , EF 2 , EF 3 , and EF 4  as described in the following specific configuration are provided at input sections of the first to third differential amplifiers  35   a ,  35   b , and  35   c.    
   In the Gilbert mixer of improved type as shown in  FIG. 5B , fifth and sixth low pass filters LPF 5  and LPF 6  as described in the following specific configuration are provided at output sections of the second and third differential amplifiers  35   b  and  35   c.    
   That is, according to the specific configuration of the Gilbert mixer of improved type as shown in  FIG. 5B , the first low pass filter LPF 1  including first and second coils L 1  and L 2  and a ninth resistor R 9  connected in series and the second low pass filter LPF 2  including third and fourth coils L 3  and L 4  and a tenth resistor R 10  are provided, respectively, between a first pair of lines + and − for transmitting a first signal V 1  branched in phase by means of the branch circuit  34  and an earth line. 
   In addition, in this Gilbert mixer of improved type, the third low pass filter LPF 3  including fifth and sixth coils L 5  and L 6  and a eleventh resistor R 11  connected in series and the fourth low pass filter LPF 4  including seventh and eighth coils L 7  and L 8  and a twelfth resistor R 12  are provided, respectively, between a second pair of lines + and − for transmitting a second signal V 2  branched in phase by means of the branch circuit  34  and the earth line. 
   In addition, this Gilbert mixer of improved type includes seventh and eighth transistors Q 7  and Q 8  each having a base input end and an emitter output end. This mixer includes the first and second emitter follower circuits EF 1  and EF 2  in which each of the base input ends of the seventh and eighth transistors Q 7  and Q 8  is connected to each of the connecting neutral points of the first and second coils L 1  and L 2  and the third and fourth coils L 3  and L 4  as output ends of the first and second low pass filters LFP 1  and LPF 2 . 
   In addition, this Gilbert mixer of improved type includes ninth and tenth transistors Q 9  and Q 10  each having a base input end and an emitter output end. This mixer includes third and fourth emitter follower circuits EF 3  and EF 4  in which each of the base input ends of the ninth and tenth transistors Q 9  and Q 10  is connected to each of the connecting neutral points of the fifth and sixth coils L 5  and L 6  and the seventh and eighth coils L 7  and L 8  as output ends of the third and fourth low pass filters LFP 3  and LPF 4 . 
   From among the first and second pairs of lines + and − for transmitting the first and second signals V 1  and V 2 , the second and third bias power sources Vb 2  and Vb 3  are connected between one line − and an earth line. 
   Here, in each of the emitters of the seventh and eighth transistors Q 7  and Q 8  and the ninth and tenth transistors Q 9  and Q 10 , thirteenth to sixteenth resistors are connected at a connecting neutral point between the constant current source I 1  and the bias power source Vb 1 , respectively. 
   In addition, each of the base input ends of the first and second transistors Q 1  and Q 2  of the first differential amplifier  35   a  is connected to each of the output ends of the first and second emitter follower circuits EF 1  and EF 2 . 
   In addition, each of the base input ends of the third and sixth transistors Q 1  and Q 2  of the second and third differential amplifiers  35   b  and  35   c  is connected to each of the output ends of the third and fourth emitter follower circuits EF 3  and EF 4 . 
   In addition, a collector output end of the third transistor Q 3  of the second differential amplifier  35   b  and a collector output end of the fifth transistor Q 5  of the third differential amplifier  35   c  are connected in common to a load resistor R 3  via the ninth coil L 9 , and are connected to the first output end OUT 1  via the tenth coil L 10 . 
   Here, the ninth coil L 9 , the load resistor R 3 , and the tenth coil L 10  configure a fifth low pass filter LPF 5 . 
   In addition, a collector output end of the fourth transistor Q 4  of the second differential amplifier  35   b  and a collector output end of the sixth transistor Q 6  of the third differential amplifier  35   c  are connected to an earth line via an eleventh coil L 11  in common and via a load resistor R 4 , and are connected to the second output end OUT 2  via a twelfth coil L 12 . 
   Here, the eleventh coil L 11 , the load resistor R 4 , and the twelfth coil L 12  configure the sixth low pass filter LPF 6 . 
   In this manner, at least one of the linearly multiplied output —(V 1 ×V 2 ) and (V 1 ×V 2 ) of the first and second signals V 1  and V 2  can be lead out from the first and second output ends OUT 1  and OUT 2 . 
   That is, a sine wave shaped signal S (t) as shown in  FIG. 6A , for example, is inputted in a burst shape in phase as first and second signals V 1  and V 2  to the thus configured improved linear multiplier  35  using the Gilbert mixer shown in  FIG. 5B , its output signal is produced as a waveform (S (t) 2 ) obtained by squaring an input signal S (t), as shown in  FIG. 6B . Its envelope (baseband) W is proportional to power of the input signal S (t) as is the case with the basic linear multiplier  35  using the Gilbert mixer shown in  FIG. 5A . 
   In addition, the linear multiplier  35  using the Gilbert mixer shown in  FIG. 5B  improved for use in the detector circuit  33  can be configured to be very small sized with a microwave monolithic integrated circuit (MMC). Moreover, there is no need for supplying a local signal unlike a conventional quadrature type detector circuit, and thus, power consumption is reduced concurrently, as is the case with the basic linear multiplier  35  using the Gilbert mixer shown in  FIG. 5A . 
   Hence, in the improved Gilbert mixer shown in  FIG. 5B  configured as described above, the first to fourth low pass filters LPF 1 , LPF 2 , LPF 3 , and LPF 4  and the first to fourth emitter follower circuits EF 1 , EF 2 , EF 3 , and EF 4  each having high Q are provided at input sections of the first to third differential amplifiers  35   a ,  35   b , and  35   c . In this manner, input impedance is enhanced, and a peaking effect is attained. 
   In addition, in the Gilbert mixer of improved type as shown in  FIG. 5B , the fifth and sixth low pass filters LPF 5  and LPF 6  are provided at output sections of the second and third differential amplifiers  35   b  and  35   c , and thereby a peaking effect is attained. 
   In this manner, the Gilbert mixer of improved type as shown in  FIG. 5B  is improved so as to enable impedance matching or peaking correction and the like at an input/output section of the linear multiplier  35  using the Gilbert mixer having a basic circuit configuration as shown in  FIG. 5A . Thus, its response characteristics are effectively improved, and the improved linear multiplier  35  using the Gilbert mixer which can be fully used in UWB can be provided. 
     FIG. 7  shows a measurement result of frequency characteristics of the linear multiplier  35  using the Gilbert mixer of improved type shown in  FIG. 5B . 
   That is, according to the measurement result of the frequency characteristics of the linear multiplier  35  using the Gilbert mixer of improved type shown in  FIG. 7 , a bandwidth within −3 dB extends to about 27 GHz, and it is determined that sufficient adaptability be provided to short range radars whose UWB center is a carrier frequency (for example, 26 GHz). 
     FIG. 8  shows a waveform (averaging number 64) in the case of observing by means of an observation oscilloscope a baseband component W obtained by applying 7 GHz bandwidth limitation to an output relevant to an input signal having a pulse width 1 ns of the linear multiplier  35  using the Gilbert mixer of improved type as shown in  FIG. 5B  by means of a low pass filter  36 . 
   That is, according to the observation waveform shown in  FIG. 8 , an average rise time obtained by a computing function of the observation oscilloscope is set to about 59 ps, and an average fall time is set to about 36 ps (a fall time from 80% to 20%), and it is found that extremely high speed response characteristics are provided. 
     FIG. 9  shows a measurement result of input/output characteristics of the linear multiplier  35  using the Gilbert mixer of improved type as shown in  FIG. 5B . 
   That is, according to a measurement result shown in  FIG. 9 , it is found that good linearity is obtained in a wide range from −30 dBm to −5 dBm in input level. 
   Therefore, the level of an input signal (V 1 , V 2 ) is controlled in the range from −30 dBm to −5 dBm, and thereby an output of the improved linear multiplier  35  using the Gilbert mixer shown in  FIG. 5B  precisely indicates power of the input signal. 
   In addition, a baseband signal W obtained by means of the detector circuit  33  as described above is inputted to a sample hold circuit  37 . 
   The sample hold circuit  37 , as its principle is shown in  FIG. 10 , has a configuration for inputting a baseband signal W via a switch  37   c  to an integrator circuit using a resistor  37   a  and a capacitor  37   b.    
   While a pulse signal Pc from a pulse generator  38  is at a high level (may be at a low level), the switch  37   c  is closed, and the baseband signal W is integrated. When the pulse signal Pc is at a low level, the switch  37   c  is opened, and an integration result is held by means of the capacitor  37   b.    
   While a description is given assuming that a sampling cycle of the sample hold circuit  37 , i.e., a cycle of the pulse signal Pc, is equal to that of a trigger signal G, the sampling cycle may be an integer multiple of a cycle Tg of the trigger signal G. 
   The pulse generator  38  receives a signal G′ synchronized with the trigger signal G (or trigger signal G itself), and delays by a time interval Td specified by the control section  50  in response to the signal G. In addition, this pulse generator generates a pulse signal Pc having a width Tc specified by the control section  50  and outputs the generated signal to the sample hold circuit  37 . 
   A signal H held after integrated by the sample hold circuit  37  is converted into a digital value by means of an A/D converter  39  immediately after being held, and the converted digital value is inputted to the signal processor section  40 . 
   The signal processor section  40  analyzes the object  1   a  which exists in the space  1  based on a signal H obtained at the receiver section  30 ; broadcasts its analysis result by an output device, although not shown (for example, display and voice generator); and notifies information required for control to the control section  50 . 
   The control section  50  makes a variety of predetermined controls with respect to at least one of the transmitter section  21  and the receiver section  30  in accordance with a schedule (program) predetermined with respect to this short range radar  20  or in response to a processing result of the signal processor section  40 . 
   Now, one example of operation of this short range radar  20  will be described here. 
   The control section  50  sets a gain of the power amplifier  25  to a predetermined value in initial setting of an investigating operation by this short range radar  20 ; sets a gain of the LNA 32  at a maximum, for example; and supplies a trigger signal G having a cycle Tg (for example, 10 μs) to the pulse generator  23  of the transmitter section  21 . 
   In this manner, when a pulse signal Pa having a width Tp (for example, 1 ns) as shown in  FIG. 11A  is inputted to the oscillator  24  of the transmitter section  21 , the transmitter section  21  radiates a short range wave Pt having a width Tp to the space  1  at a carrier frequency Fc (for example, 26 GHz) as shown in  FIG. 11B  from the transmitter antenna  22  via the power amplifier  25  and BRF  26 . 
   At this time, power supply to the transmitter section  21  is provided to only an output period of the short range wave Pt (or very limited period including the output period) by means of the control section  50 . 
   In this manner, a time interval at which power is supplied to the transmitter section  21  is substantially 1/10000 of the whole cycle Tg, and thus, wasteful power consumption does not occur. 
   The short range wave Pt radiated from the transmitter section  21  is reflected by the object  1   a  which exists in the space  1 , and the reflection wave Pr is received by means of the receiver antenna  31  of the receiver section  30  after being delayed by a time interval Tx corresponding to a reciprocal distance from a transmission timing of each short range wave Pt to the object  1   a , as shown in  FIG. 11C , for example. 
   In the receiver section  30 , after the signal R of the thus received reflection wave Pr has been amplified by means of the LNA 32 , the amplified signal is subjected to bandwidth limitation by means of the BPF  41 , and noise power is reduced. In addition, after the signal R′ of the reflection wave Pr outputted from the BPF  41  has been branched into two sections, the first signal V 1  and second signal V 2  in phase by means of the branch circuit  34  of the detector circuit  33 , the branched signals are detected by means of the linear multiplier  35  and the low pass filter  36 , thereby detecting a baseband component W as shown in  FIG. 11D . 
   On the other hand, in the sample hold circuit  37 , a pulse signal Pc having a width (for example, 1 ns) as shown in  FIG. 11E  is inputted to be delayed by Td, 2Td, 3Td, . . . and nTd (n is an integer) from each transmission timing of the short range wave Pt. 
   Here, a description will be given with respect to a case in which the delay time Td is equal to a width of the pulse Pc. 
   In addition, assuming that a distance up to a distal end of the space  1  targeted for investigation is within 15 m, a time for a radio wave to reciprocate the distance of 15 m is substantially 100 ns. 
   Therefore, by delaying a transmission timing of a short range wave Pt by a maximum of 100 Td, as long as the reflection waves Pr is within the range of 15 m, these reflection waves Pr can be fully included in coverage. 
   As shown in  FIGS. 11C ,  11 D, and  11 E, the first to third pulse signals Pc do not overlap a baseband component W, and thus, the sample hold circuit  37  integrates only a noise component, and its integration result and hold value are substantially zero. 
   When fourth and fifth pulse signals Pc overlap a baseband component W, as shown in  FIG. 11F , the baseband signal W is integrated within a high level period of the pulse signals Pc, and the integration results H 1  and H 2  are held. In this manner, the hold values H 1  and H 2  are converted into digital values by means of the A/D converter  39 , and the converted digital values are outputted to the signal processor section  40  in a manner as shown in  FIG. 11G . 
   The signal processor section  40  detects a distance up to the object  1   a  and the object size based on these hold values H 1  and H 2 . 
   That is, when a hold value H equal to or greater than a predetermined level has been inputted, for example, the signal processor section  40  detects a distance up to the object  1   a  according to how many samplings have been performed before the input is obtained. 
   In addition, in the case where a hold value H equal to or greater than a predetermined level is continuous, the signal processor section  40  detects the size of the object  1   a  according to its continuous number. 
   This detection information is notified to the control section  50 . 
   When the detection information notified from the signal processor section  40  indicates that a distance up to the object  1   a  is short, and the intensity of the reflection wave Pr is high, the control section  50  reduces a gain of the LNA 2  of the receiver section  30  so that an input level of the detector circuit  33  is within the range of linear operation of the linear multiplier  35 . 
   In this case, the control section  50  controls a gain of the power amplifier  25  of the transmitter section  21  to be reduced if necessary. 
   In this manner, during next investigation, a more precise baseband component W is detected in the detector circuit  33  of the receiver  30 . 
   In addition, in the case where the detection information notified from the signal processor section  40  indicates that there is a need for analyzing a weak reflection wave Pr from the vicinity of a distal end of an investigation space  1 , the control section  50  controls a gain of the power amplifier  25  of the transmitter section  21  to be increased. 
   In this manner, during next investigation, a more precise baseband component W is detected in the detector circuit  33  of the receiver section  30 . 
   In addition, the control section  50  makes control so as to obtain necessary investigation information by appropriately varying the integration time Tc of the sample hold circuit  37  according to the state of the investigation space  1 , the size of the object  1   a  and the like. 
   In this case, although the control section  50  makes control for stopping power supply excluding only a period in which a short range wave Pt is radiated with respect to the transmitter section  21 , this control section does not make such control with respect to the receiver section  30  at all. 
   Hence, as described previously, in the case where a time interval corresponding to the investigation range is 100 ns, and the radiation cycle Tg of the short range wave Pt is 10 μs, in fact, only about 1/100 in that cycle Tg is utilized. 
   Therefore, during the remaining period (that is, about 99/100 in the cycle Tg), power supply to the receiver section  30  is stopped by the control section  50 , and thereby power consumption can be further reduced. 
   In addition, for example, in the case where a hold output H equal to or greater than a predetermined level cannot be obtained by radiation of 100 short range waves Pt, the signal processor section  40  judges that no object becomes an obstacle in the investigation range, and notifies the fact to the control section  50 . 
   The control section  50  having received this notification stops power supply to the transmitter section  21  and the receiver section  30  for a predetermined period (for example, 1 ms); restarts power supply after elapse of the predetermined time; and makes control for repeating the investigating operation as described above. 
   Power consumption of the whole short range radars can be remarkably reduced and battery can be driven by controlling power supply to the transmitter section  21  and the receiver section  30  by means of the control section  50 . 
   In this manner, it becomes possible to provide portable short range radars. 
   In the foregoing description, in the sample hold circuit  37 , investigation is made while a integration timing is shifted in a short integration time. 
   Hence, for example, at the initial stage of investigation, an integration time is set at a time interval (for example, 100 ns) corresponding to an investigation distance (that is, is set to a full range), thereby making it possible to speedily grasp the presence or absence of an object by one short pulse radiation. 
   Second Embodiment 
     FIG. 12  is a block diagram depicting a configuration of essential portions of a second embodiment of short range radars according to the present invention. 
   As described above, in the integration type sample hold circuit  37  according to the first embodiment, an electric discharge due to a leakage occurs, thus making it difficult to hold a voltage for a long period of time. 
   In such a case, as shown in  FIG. 12 , a plurality of sample hold circuits, in this example, four sample hold circuits  37 A,  37 B,  37 C, and  37 D and four A/D converters  39 A,  39 B,  39 C, and  39 D are provided in parallel. 
   In addition, for example, Pc (t), Pc (t+Te/4), Pc (t+Te/2), and Pc (t+3Te/4) may be applied from a pulse generator  38 ′ as a plurality of pulse signals whose generation times are different from each other so that the sample hold circuits  37 A,  37 B,  37 C, and  37 D each carry out integration at their respectively different periods with respect to an output signal W of the detector circuit  33 . 
   Namely, with respect to the above example of numeric values, the whole integration time Te is 100 ns, and the pulse generator  38 ′ provides four pulse signals Pc (t), Pc (t+25 ns), Pc (t+50 ns), and Pc (t+75 ns) whose width is 25 ns (=Te/4) and each of which is delayed by 25 ns (=Te/4) to each of the sample holds circuits  37 A,  37 B,  37 C, and  37 D. 
   Then, hold values Ha, Hb, Hc, and Hd from the sample hold circuits  37 A,  37 B,  37 C, and  37 D may be outputted to the signal processor section  40  after converted into digital values by means of the A/D converters  39 A,  39 B,  39 C, and  39 D, respectively. 
   In this case, the signal processor section  40  analyzes whether or not an object  1   a  exists in an investigation space  1  based on at least one of the four hold values Ha, Hb, Hc, and Hd from the sample hold circuits  37 A,  37 B,  37 C, and  37 D. 
   Even if the first three hold values Ha, Hb, and Hc cannot be discriminated from among the four hold values Ha, Hb, Hc, and Hd due to electric discharge caused by a leakage, during this analysis, the signal processor section  40  can analyze whether or not the object  1   a  exists in the investigating space  1  based on the immediately following fourth hold value Hd. 
   Third Embodiment 
     FIG. 13  is a block diagram depicting a configuration of essential portions of a third embodiment of short range radars according to the present invention. 
   In  FIG. 13 , the same constituent elements having a configuration of short range radars according to the first embodiment shown in  FIG. 1  are designated by the same reference numerals. A duplicate description is omitted here. 
   As described above, in the short range radars according to the present invention, a linear multiplier  35  is used for a detector circuit  33 , and thereby there is no need for using a local signal unlike a conventional quadrature type detector circuit for use in pulse radars. Thus, a short pulse radar  20 ′ in a diversity system as shown in  FIG. 13  can be provided very easily. 
   In this short pulse radar  20 ′, two pairs of receiver sections  30 A and  30 B and two pairs of A/D converters  39 A and  39 B allocated in a state in which respective receiving antennas  31   a  and  31   b  are spaced from each other by a predetermined distance are provided with respect to one transmitter section  21 , one signal processor section  40 , and one control section  50 . 
   Then, with respect to signals of two reflection waves Pr and Pr′ reflected in different directions from an object  1   a , the receiver sections  30 A and  30 B each apply an detecting process using a linear multiplier  35  and an integrating process using a sample hold circuit  37  in the same manner as in the receiver section  30  according to the first embodiment shown in  FIG. 1 . In addition, these two outputs Ha and Hb are converted into digital signals by means of each of the A/D converters  39 A and  39 B, and then, a delay time difference between the two reflection waves Pr and Pr′ is detected by means of the signal processor section  40 , thereby making it possible to grasp a direction, a moving direction and the like of the object  1   a.    
   Thus, even in the case where a plurality of receiver sections  30 A,  30 A are provided, there is no need for local signal cable run or shielding like the receiver section  30  according to the first embodiment shown in  FIG. 1 . In addition, detection can be carried out by means of the detector circuit  33  which includes independent linear multipliers  35 , respectively, and thus, equipment designing of short range radars becomes very easy. 
   Therefore, as described above, according to the present invention, there can be provided short range radars and control method thereof which solve the problems associated with the conventional technique, and which is small in size and low in power consumption so as to be available in UWB.