Patent Publication Number: US-8541991-B2

Title: Power converter with controller operable in selected modes of operation

Description:
This application is a continuation-in-part of patent application Ser. No. 12/707,420, entitled “Power Converter with Power Switch Operable in Controlled Current Mode,” filed on Feb. 17, 2010 now U.S. Pat. No. 8,154,261, which is a continuation of patent application Ser. No. 12/103,993, entitled “Power Converter with Power Switch Operable in Controlled Current Mode,” filed on Apr. 16, 2008 (now, U.S. Pat. No. 7,679,342), which applications are incorporated herein by reference. 
    
    
     TECHNICAL FIELD 
     The present invention is directed, in general, to power electronics and, more specifically, to a power converter and method of controlling the same for selected modes of operation. 
     BACKGROUND 
     A switch-mode power converter (also referred to as a “power converter” or “regulator”) is a power supply or power processing circuit that converts an input voltage waveform into a specified output voltage waveform. DC-DC power converters convert a direct current (“dc”) input voltage into a dc output voltage. Controllers associated with the power converters manage an operation thereof by controlling the conduction periods of power switches employed therein. Generally, the controllers are coupled between an input and output of the power converter in a feedback loop configuration (also referred to as a “control loop” or “closed control loop”). 
     Typically, the controller measures an output characteristic (e.g., an output voltage, an output current, or a combination of an output voltage and an output current) of the power converter, and based thereon modifies a duty cycle of the power switches of the power converter. The duty cycle is a ratio represented by a conduction period of a power switch to a switching period thereof. Thus, if a power switch conducts for half of the switching period, the duty cycle for the power switch would be 0.5 (or 50 percent (“%”)). Additionally, as voltage or current for systems, such as a microprocessor powered by the power converter, dynamically change (e.g., as a computational load on the microprocessor changes), the controller should be configured to dynamically increase or decrease the duty cycle of the power switches therein to maintain an output characteristic such as an output voltage at a desired value. 
     In an exemplary application, the power converters have the capability to convert an unregulated input voltage, such as five volts, supplied by an input voltage source to a lower, regulated, output voltage, such as 2.5 volts, to power a load. To provide the voltage conversion and regulation functions, the power converters include active power switches such as metal-oxide semiconductor field-effect transistors (“MOSFETs”) that are coupled to the voltage source and periodically switch a reactive circuit element such as an inductor to the voltage source at a switching frequency that may be on the order of five megahertz. 
     In typical applications of dc-dc power converters, power conversion efficiency is an important parameter that directly affects the physical size of the end product, the cost and market acceptance. Active power switches that are either fully on with low forward voltage drop or fully off with minimal leakage current provide a recognized advantage for power conversion efficiency in comparison with previous designs that utilized a dissipative “pass” transistor to regulate an output characteristic or a passive diode to provide a rectification function. Previous designs using pass transistors and passive diodes produced operating power conversion efficiencies of roughly 40-70% in many applications. The use of active power switches in many recent power converter designs, particularly as synchronous rectifiers for low output voltages, has increased operating efficiency at full rated load to 90% or more. 
     A continuing problem with power converters is preserving power conversion efficiency at low levels of output current. Low efficiency at low output currents is a result of power inherently lost by parasitic elements in the power switches and by losses induced in reactive circuit elements, particularly inductors coupled to the active power switches. Further losses are also generated in the control and drive circuits coupled to the active power switches. Ultimately, as the output current of a power converter approaches zero, the fixed losses in the power switches, the inductive circuit elements, and the control circuits cause power conversion efficiency also to approach zero. 
     Various approaches are known to improve power conversion efficiency at low output currents. One approach used with resonant power conversion topologies reduces switching frequency of active power switches for low output current. Another approach, as described by X. Zhou, et al., in the paper entitled “Improved Light-Load Efficiency for Synchronous Rectifier Voltage Regulation Module,” IEEE Transactions on Power Electronics, Volume 15, Number 5, September 2000, pp. 826-834, which is incorporated herein by reference, utilizes duty cycle adjustments to adjust switching frequency or to disable a synchronous rectifier switch. A further approach, as described by M. E. Wilcox, et al. (“Wilcox”), in U.S. Pat. No. 6,580,258, entitled “Control Circuit and Method for Maintaining High Efficiency Over Broad Current Ranges in a Switching Regulator Circuit,” issued Jun. 17, 2003, which is incorporated herein by reference, generates a control signal to intermittently turn off one or more active power switches under light load operating conditions when the output voltage of the power converter can be maintained at a regulated voltage by the charge on an output capacitor. Of course, when an output voltage from a power converter is temporarily discontinued, such as when the load coupled thereto is not performing an active function, the power converter can be disabled by an enable/disable signal, generated either at a system or manual level, which is a process commonly used, even in quite early power converter designs. 
     However, resonant power conversion topologies are frequently a poor choice in many applications due to an inherently disadvantageous waveform structure in resonant circuits and the resulting inefficient use of semiconductor power switches to execute the resonant power conversion process at high levels of load current. Intermittently turning off one or more active power switches under light load operating conditions as described by Wilcox still generates associated switching losses when the active power switches are periodically operated to maintain charge on an output filter capacitor. Thus, the problem of providing high power conversion efficiency at light load currents still remains an unresolved issue. 
     Accordingly, what is needed in the art is a power converter and related method to provide high power conversion efficiency in a switch-mode power converter, especially at light load currents, that overcomes deficiencies in the prior art. 
     SUMMARY OF THE INVENTION 
     These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by advantageous embodiments of the present invention, including a power converter and method of controlling the same for selected modes of operation. In one embodiment, the power converter includes a first power switch coupled to a source of electrical power and a second power switch coupled to the first power switch and to an output terminal of the power converter. The power converter also includes a controller configured to control an operation of the first and second power switches during selected modes of operation. 
     The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures or processes for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  illustrates a diagram of an embodiment of a power converter constructed according to the principles of the present invention; 
         FIG. 2  illustrates a diagram of portions of the power converter illustrated in  FIG. 1  constructed according to the principles of the present invention; 
         FIG. 3  illustrates a waveform diagram of an exemplary operation associated with a power switch of a power converter in accordance with the principles of the present invention; 
         FIG. 4  illustrates a diagram of an embodiment of portions of a power converter constructed according to the principles of the present invention; 
         FIGS. 5A to 5D  illustrate waveform diagrams of exemplary operations of a power converter in accordance with the principles of the present invention; 
         FIGS. 6A and 6B  illustrate waveform diagrams of exemplary operations associated with a power switch of a power converter in accordance with the principles of the present invention; 
         FIGS. 7 and 8  illustrate flow diagrams of embodiments of methods of operating a controller in accordance with the principles of the present invention; 
         FIGS. 9A and 9B  illustrate waveform diagrams of exemplary operations associated with power switches of a power converter in accordance with the principles of the present invention; 
         FIGS. 10 and 11  illustrate flow diagrams of embodiments of methods of operating a controller in accordance with the principles of the present invention; 
         FIGS. 12 to 14  illustrate schematic drawings of embodiments of portions of power converters constructed according to the principles of the present invention; and 
         FIGS. 15 and 16  illustrate schematic drawings of embodiments of portions of controllers constructed according to the principles of the present invention. 
     
    
    
     Corresponding numerals and symbols in the different figures generally refer to corresponding parts unless otherwise indicated, and may not be redescribed in the interest of brevity after the first instance. The FIGUREs are drawn to clearly illustrate the relevant aspects of exemplary embodiments. 
     DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
     The making and using of the presently exemplary embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention. 
     The present invention will be described with respect to exemplary embodiments in a specific context, namely, a power converter including a controller responsive to a level of output current or other parameters to regulate an output characteristic and methods of operating the same. While the principles of the present invention will be described in the environment of a power converter, any application that may benefit from a power converter, such as a power amplifier, including a controller responsive to a level of current to regulate an output characteristic is well within the broad scope of the present invention. 
     Referring initially to  FIG. 1 , illustrated is a diagram of an embodiment of a power converter constructed according to the principles of the present invention. The power converter includes a power train  110 , a controller  120 , and a driver (e.g., a gate driver)  130 , and provides power to a system/load such as a microprocessor (not shown) coupled to output terminals  140 ,  141 . The controller  120  is responsive to a level of output current I out  to regulate an output characteristic of the power converter. While in the illustrated embodiment the power train  110  employs a buck converter topology, those skilled in the art should understand that other converter topologies such as a forward converter topology are well within the broad scope of the present invention. 
     The power train  110  of the power converter receives an input voltage V in  from a source of electrical power (represented by battery  150 ) at an input thereof and provides a regulated output voltage V out  at the output terminals  140 ,  141 , or other output characteristic such as an output current I out . In keeping with the principles of a buck converter topology, the output voltage V out  is generally less than the input voltage V in  such that a switching operation of the power converter can regulate the output voltage V out . 
     In a first mode of operation (also referred to as a pulse-width modulation (“PWM”) mode of operation), wherein substantial output current I out  is delivered to the output terminals  140 ,  141 , a main power switch Q mn  [e.g., a p-channel metal oxide semiconductor field effect transistor (“MOSFET”) embodied in a p-type laterally diffused metal oxide semiconductor (“P-LDMOS”) device], is enabled to conduct in response to a gate drive signal S DRV1  for a primary interval (generally co-existent with a primary duty cycle “D” of a switching cycle) and couples the input voltage V in  to an output filter inductor (or output inductor) L out . During the primary interval, an inductor current I Lout  flowing through the output inductor L out  increases as current flows from the input to the output of the power train  110 . An ac component of the inductor current I Lout  is filtered by an output capacitor C out  to provide the output current I out  at an output of the power converter. The power converter generally operates in the PWM mode of operation, for example and without limitation, when the output current I out  is of sufficient magnitude that the inductor current I Lout  in the output inductor L out  or in a power switch does not reverse direction. 
     During a complementary interval during the PWM mode of operation (generally co-existent with a complementary duty cycle “1-D” of the switching cycle), the main power switch Q mn  is transitioned to a non-conducting state and an auxiliary power switch Q aux  [e.g., an n-channel MOSFET embodied in an n-type laterally diffused metal oxide semiconductor (“N-LDMOS”) device], coupled to the output inductor L out , is enabled to conduct in response to a gate drive signal S DRV2 . The auxiliary power switch Q aux  provides a path to maintain a continuity of the inductor current I Lout  flowing through the output inductor L out . During the complementary interval, the inductor current I Lout  flowing through the output inductor L out  decreases. In general, during the PWM mode of operation, the duty cycle of the main and auxiliary power switches Q mn , Q aux  may be adjusted to maintain a regulation of the output voltage V out  of the power converter. Those skilled in the art should understand, however, that the conduction periods for the main and auxiliary power switches Q mn , Q aux  may be separated by a small time interval to avoid cross conduction therebetween and beneficially to reduce the switching losses associated with the power converter. Those skilled in the art should understand further that terms used herein such as “current reversal” or a reference to a particular level of a physical quantity such as “zero current” are to be understood within the context of a physical apparatus with attendant and practical accuracy limitations. For example, one cannot know or measure the precise instant that a current that reverses direction passes through a current level of zero. 
     The controller  120  of the power converter receives an output characteristic (e.g., the output current I out  and/or the output voltage V out ) of the power converter, and a desired output characteristic such as a desired system voltage V system  from an internal source or from an external source that may be associated with the load. In an advantageous embodiment, the controller  120  may be coupled to a current sensor, such as current sensor  160  to sense a power converter current such as an inductor current I Lout  or the output current I out . In a further advantageous embodiment, the controller  120  may be coupled to a current sensor to sense a current in a power switch. Thus, a current sensor may be employed by controller  120  to select the PWM mode of operation of the power converter by comparing a sensed current with a fixed or adjustable current threshold level. 
     The controller  120  may also be coupled to an input characteristic (e.g., the input voltage V in ) of the power converter and to a return lead of the source of electrical power (again, represented by battery  150 ) as illustrated in  FIG. 1  to provide a ground connection therefor. While only a single ground connection is illustrated in the present embodiment, those skilled in the art should understand that multiple ground connections may be employed for use within the controller  120 . A decoupling capacitor C dec  may be coupled as illustrated in the FIGURE to the path from the input voltage V in  to the controller  120 . The decoupling capacitor C dec  is generally configured to absorb high frequency noise signals associated with the source of electrical power to protect the controller  120 . 
     In accordance with the aforementioned characteristics, during the PWM mode of operation, the controller  120  provides a signal (e.g., a pulse-width modulated signal S PWM ) to control the duty cycle and a frequency of the main and auxiliary power switches Q mn , Q aux  of the power train  110  to regulate the output voltage V out  or other output characteristic thereof. The controller  120  in some applications may also provide a complement of the pulse-width modulated signal S PWM  during the PWM mode of operation (e.g., a complementary pulse-width modulated signal S 1-PWM ) in accordance with the aforementioned characteristics. Any controller adapted to control at least one power switch of the power converter is well within the broad scope of the present invention. As an example, a controller employing digital circuitry is disclosed in U.S. Pat. No. 7,038,438, entitled “Controller for a Power Converter and a Method of Controlling a Switch Thereof,” issued May 2, 2006, to Dwarakanath, et al., and U.S. Pat. No. 7,019,505, entitled “Digital Controller for a Power Converter Employing Selectable Phases of a Clock Signal,” issued Mar. 28, 2006, to Dwarakanath, et al., which are incorporated herein by reference. 
     The power converter also includes a driver (e.g., a gate driver)  130  to provide gate drive signals S DRV1 , S DRV2  to control conductivity of the main and auxiliary power switches Q mn , Q aux , respectively, responsive to the pulse-width modulated signal S PWM  (and, if necessary, the control the complementary pulse-width modulated signal S 1-PWM ) provided by the controller  120 . There are a number of viable alternatives to implement a driver  130  that include techniques to provide sufficient signal delays to prevent crosscurrents when controlling multiple power switches in the power converter. The driver  130  typically includes switching circuitry incorporating a plurality of driver switches that cooperate to provide the gate drive signals S DRV1 , S DRV2  to the main and auxiliary power switches Q mn , Q aux . Of course, any driver  130  capable of providing the gate drive signals S DRV1 , S DRV2  to control a power switch is well within the broad scope of the present invention. As an example, a driver is disclosed in U.S. Pat. No. 7,330,017, entitled “Driver for a Power Converter and a Method of Driving a Switch Thereof,” issued Feb. 12, 2008, to Dwarakanath, et al., and a power switch is disclosed in U.S. Pat. No. 7,230,302, entitled “Laterally Diffused Metal Oxide Semiconductor Device and Method of Forming the Same,” issued Jun. 12, 2007, to Lotfi, et al., and in U.S. Pat. No. 7,214,985, entitled “Integrated Circuit Incorporating Higher Voltage Devices and Low Voltage Devices Therein,” issued May 8, 2007, to Lotfi, et al., which are incorporated herein by reference. 
     According to the principles of the present invention, the main and auxiliary power switches Q mn , Q aux  are typically power switches that can be incorporated into a semiconductor device in an integrated circuit proximate control or signal processing devices that perform many of the control functions of the controller  120  of the power converter. The control and signal processing devices are typically complementary metal-oxide semiconductor (“CMOS”) devices such as p-type metal oxide semiconductor (“PMOS”) devices and n-type metal oxide semiconductor (“NMOS”) devices. The PMOS and NMOS devices may also be referred to as p-channel and n-channel MOSFETs, respectively. 
     In a switch-mode power converter, such as the buck power converter illustrated and described with reference to  FIG. 1 , the duty cycle of a power switch, such as the main power switch Q mn  previously described herein, determines the steady-state ratio of a power converter output voltage V out  to its input voltage V in . In particular, for a buck power converter typology operating in a continuous conduction mode (“CCM”), duty cycle determines the ratio of output voltage to input voltage (ignoring certain losses within the power converter) according to the equation:
 
 D=V   out   /V   in .  (1)
 
     In an alternative power converter typology, such as a boost topology, duty cycle determines the ratio of output to input voltage (again, ignoring certain losses within the power converter) operating in a continuous conduction mode according to the equation:
 
 D=V   in   /V   out .  (2)
 
This reciprocal relationship for the ratio of input and output voltages of buck and boost topologies recognizes that a buck power converter topology employing synchronous rectifiers is operable as a boost topology with its input and output reversed, and vice versa. Other switch-mode power converter topologies such as a buck-boost, forward, Cúk, etc., are characterized by further relationships, well known in the art, for a ratio of output voltage to input voltage, for a particular operating condition such as continuous conduction mode.
 
     A controller, such as controller  120  illustrated in  FIG. 1 , typically regulates an output characteristic of a power converter by controlling a duty cycle of a power switch. Duty cycle is generally controlled by comparing a sawtooth voltage waveform with a controlled threshold voltage produced by an error amplifier configured to sense an output voltage or other output characteristic. 
     A load coupled to a power converter may sometimes operate for a period of time in an idle mode wherein the load draws a relatively small but non-zero current from the power converter, for example, less than one percent of its normal load current. Under such operating conditions, wherein power conversion efficiency of the power converter is typically very low, it is preferable to provide high power conversion efficiency, particularly when the power converter is powered from a portable energy source such as a battery. 
     Referring to  FIG. 2 , illustrated is a diagram of portions of the power converter illustrated in  FIG. 1  constructed according to the principles of the present invention. A switch-current control subsystem  200  of the power converter is configured to control a current flowing through a main power switch Q mn  during the complementary duty cycle 1-D of a switching cycle. The switch-current control subsystem  200  includes a gate driver  201  for the main power switch Q mn . The gate driver  201  includes driver switches such as a p-channel field-effect transistor (“FET”)  224  and an n-channel FET  222 , with their gates coupled together and driven by a pulse-width modulated signal S PWM . The pulse-width modulated signal S PWM  may be created by a controller such as the controller  120  illustrated and described with respect to  FIG. 1 . The source, gate, and drain of the respective driver switches are labeled with “s,” “g,” and “d,” respectively. 
     Producing a current flowing through the main power switch Q mn  during the complementary duty cycle 1-D of a switching cycle when a power converter is lightly loaded, such as producing less than five percent of its rated output power, can provide substantial efficiency improvement over a switching mode wherein both the main and auxiliary power switches Q mn , Q aux  are continuously enabled to conduct in complementary intervals of time. A switching mode wherein the main and auxiliary power switches Q mn , Q aux  are enabled to conduct in complementary intervals of time can produce substantial bidirectional current through reactive circuit elements, such as an output inductor L out , contributing thereby substantial power losses. In addition, producing a light current flowing through the main power switch Q mn  during the complementary duty cycle 1-D of a switching cycle when a power converter is lightly loaded can be a more economical alternative than providing a large output capacitance to maintain an output voltage V out  of a power converter during a mode of operation wherein all switching is temporarily disabled. 
     The switch-current control subsystem  200  illustrated in  FIG. 2  is configured to control a gate voltage for the main power switch Q mn  when it would ordinarily be disabled to conduct (i.e., during the complementary duty cycle 1-D of a switching cycle). The gate voltage of the main power switch Q mn  is controlled during the complementary duty cycle 1-D of a switching cycle so that the main power switch Q mn  can conduct a controlled, light current to the load (i.e., an output current I out  of the power converter). 
     When the pulse-width modulated signal S PWM  is high, the gate driver  201  couples the gate of the main power switch Q mn  to ground, turning it on. When pulse-width modulated signal S PWM  is low, the gate driver  201  couples the gate of the main power switch Q mn  to the output of an operational amplifier  214 . The output of the operational amplifier  214  is controlled to enable the main power switch Q mn  to conduct a controlled, light current (e.g., a controlled current level such as a remnant current level) when the pulse-width modulated signal S PWM  is low. 
     A p-channel FET  212  with its gate coupled to its drain operates as a diode in forward conduction. The drain current of the p-channel FET  212  is controlled by a resistor  216 , which is coupled substantially across the input voltage V in  (less the diode drop of the p-channel FET  212 ). Thus, the gate of the p-channel FET  212  is set to the voltage with respect to its source that is necessary to conduct the current flowing through the resistor  216 . This gate voltage is sensed with the operational amplifier  214  and coupled to the gate of the main power switch Q mn  during the complementary duty cycle 1-D of a switching cycle. Thus, the p-channel FET  212  and the main power switch Q mn  operate as a current mirror during the complementary duty cycle 1-D of a switching cycle, wherein a current controlled by the resistor  216  and scaled by a die geometric ratio of the main power switch Q mn  to the p-channel FET  212  flows through the main power switch Q mn . Preferably, the p-channel FET  212  and the main power switch Q mn  are produced in a common manufacturing process and are configured to operate at substantially the same die temperature. Preferably, the p-channel FET  212  is a downscaled replica of the main power switch Q mn . The operation of current mirrors is well known in the art, and will not be described further in the interest of brevity. 
     Turning now to  FIG. 3 , illustrated is a waveform diagram of an exemplary operation associated with a power switch of a power converter in accordance with the principles of the present invention. In particular, the waveform represents a source-to-drain current I Qmnsd  flowing through a power switch (e.g., the main power switch Q mn  of  FIG. 2 ) wherein the output current is of sufficient magnitude that current in an output inductor or in a semiconductor power switch does not reverse direction. During the primary duty cycle D of the switching cycle, the source-to-drain current I Qmnsd  flowing through the main power switch Q mn  increases substantially linearly due to the voltage applied across an output inductor (e.g., the output inductor L out  of  FIG. 2 ). During the complementary duty cycle 1-D of the switching cycle, the source-to-drain current I Qmnsd  maintains a controlled, substantially constant current level (e.g., a remnant current level) I CC  flowing through the main power switch Q mn  that is controlled by a controller such as the switch-current control subsystem  200  of  FIG. 2 . 
     Turning now to  FIG. 4 , illustrated is a diagram of an embodiment of portions of a power converter constructed according to the principles of the present invention. A controller  400  of the power converter is operable in different modes of operation. The controller  400  regulates an output characteristic of the power converter, and is configured to control current (e.g., a controlled or remnant current level I CC ) through a main power switch Q mn  in response to a sensed or estimated power converter current. The controller  400  is advantageously operable to provide a mode of operation wherein improved power conversion efficiency is achieved at light load. Additionally, the power converter may experience an improvement in dynamic response because of an additional bias available to feed an output thereof that in turn produces a smaller decay of an output characteristic (e.g., an output voltage V out ) since the output voltage V out  is not supplied only by an output capacitor C out . The power converter includes an error amplifier  402  that senses the output voltage V out  as an output characteristic to provide a feedback signal. 
     The primary and complementary duty cycles D, 1-D of a switching cycle are established by a comparator  414  that produces a pulse-width modulated signal S PWM . The noninverting input of the comparator  414  is coupled to an error amplifier signal V EA  of the error amplifier  402 . The inverting input of the comparator  414  is coupled to a sawtooth waveform signal V sawtooth  that has a substantial positive voltage offset for the waveform valleys. 
     In a first or PWM mode of operation wherein a substantial current (an output current I out ) is delivered to a load (not shown) coupled to output terminals  440 ,  441 , the main and auxiliary power switches Q mn , Q aux  are alternately enabled to conduct, respectively, during a primary duty cycle D and a complimentary duty cycle 1-D. The primary duty cycle D and the complementary duty cycle 1-D are controlled to regulate an output characteristic of the power converter. During the PWM mode of operation, the load current (i.e., an output current I out  of the power converter) is of sufficient magnitude so that an inductor current I Lout  flowing through output inductor L out  does not reverse direction. During the complementary duty cycle 1-D, the main power switch Q mn  conducts a current controlled (e.g., a controlled current level I CC ) by a current mirror including a gate driver  401 , an operational amplifier  404 , a p-channel FET  408  and a resistor  416 . The current mirror is operable in a manner similar to that described with reference to  FIG. 2 . However, in the circuit illustrated in  FIG. 4 , the current through the main power switch Q mn  is controlled by current flowing through the resistor  416 . The resistor  416  is coupled to the error amplifier signal V EA  via an inverter  418 . The inverter  418  amplifies the output of the error amplifier  402  with gain −k. Thus, as the voltage of the error amplifier signal V EA  is reduced, the current flowing through the resistor  416  is also reduced. Correspondingly, as the voltage of the error amplifier signal V EA  is reduced, the current flowing through the main power switch Q mn  during the complementary duty cycle 1-D of the switching cycle is also reduced. 
     The error amplifier  402  senses the output voltage V out . The error amplifier  402  includes an operational amplifier  409  that includes feedback networks  405 ,  406 . The feedback networks  405 ,  406  include a parallel arrangement of a capacitor and a resistor. The error amplifier  402  further includes input networks including resistors  410 ,  412 . In a preferred embodiment, the values of components in feedback networks  405 ,  406  are equal, and the values of the resistors  410 ,  412  in the input networks are equal. The selection of component values for an error amplifier to produce a stable response of a power converter in a particular application is well known in the art, and will not be described further in the interest of brevity. The error amplifier  402  generates the error amplifier signal V EA  in response to the sensed output voltage V out  of the power converter and a desired system voltage V system . Of course, different arrangements of feedback and input networks to meet the needs of a particular application including a voltage divider coupled across output terminals  440 ,  441  are well within the broad scope of the invention. Thus, in the PWM mode of operation, the controller  400  enables alternating conduction of the main and auxiliary power switches Q mn , Q aux  while enabling a controlled current (e.g., a controlled current level I CC ) flowing through the main power switch Q mn  during the complementary duty cycle 1-D. The controlled current flowing through the main power switch Q mn  is responsive to the error amplifier signal V EA  produced by the error amplifier  402 . 
     In a second mode operation (also referred to as a hybrid mode of operation), the power converter output current I out  is insufficient to sustain unidirectional current flow in the output inductor L out  as described with respect to the PWM mode of operation. Bidirectional current flow through output inductor L out  is prevented in the power converter illustrated in  FIG. 4  by sensing the inductor current I Lout  with a current sensor  460 . The sensed inductor current I Lout  is amplified with a transresistance amplifier that includes operational amplifier  426  coupled to feedback resistor  428 . The gain of the transresistance amplifier is substantially the resistance of the resistor  428 . The output of the transresistance amplifier is coupled to an input of an AND gate  420 . 
     The output of the comparator  414  is coupled to a signal inverter  422 . The signal inverter  422  produces a high output signal during the complementary duty cycle 1-D of the switching cycle. The output of the signal inverter  422  is coupled to the other input of the AND gate  420 . Thus, the AND gate  420  produces a high gate drive signal S DRV2  for the auxiliary power switch Q aux  to enable conduction therein during the complementary duty cycle 1-D of the switching cycle, when positive inductor current I Lout  flows through the output inductor L out . In accordance therewith, the polarity of the current sensor  460  is selected, as would be known by one with ordinary skill in the art, to produce a sense of the signal at the output of operational amplifier  426  to enable conduction in the auxiliary power switch Q aux  during the complementary duty cycle 1-D of the switching cycle, when positive current flows through the output inductor L out . The current mirror previously described continues to provide a gate drive signal S DRV1  for the main power switch Q mn  to enable a current controlled by the error amplifier  402  to flow therethrough during the complementary duty cycle 1-D of the switching cycle. In this manner, during the hybrid mode of operation, a controlled current (e.g., a controlled current level I CC ) is enabled to flow through the main power switch Q mn  when the auxiliary power switch Q aux  is disabled to conduct during the complementary duty cycle 1-D of the switching cycle. The controller, therefore, is configured to control a level of current (again, e.g., a controlled current level I CC ) in the main power switch Q mn  when the auxiliary power switch Q aux  is substantially disabled to conduct (e.g., during the complementary duty cycle 1-D of the switching cycle). 
     In a third mode operation (also referred to as a light current mode of operation), the output current I out  is further reduced so that a source-to-drain current that flows through the main power switch Q mn  is controlled to the controlled current level I CC  to maintain sufficient current to power the load. As in the PWM and hybrid modes of operation, the error amplifier  402  can be used to control current flowing to the output terminal  440 , thereby regulating an output characteristic of the power converter. In the light current mode of operation, the alternately enabled conduction of the main and auxiliary power switches Q mn , Q aux  is disabled, since the source-to-drain current that flows through the main power switch Q mn  is sufficient to power the load. 
     Turning now to  FIGS. 5A to 5D , illustrated are waveform diagrams of exemplary operations of a power converter in accordance with the principles of the present invention. In the interest of maintaining continuity, the waveform diagrams will be described in part with reference to signals and components illustrated and described with respect to  FIG. 4 . More particularly, a left portion of  FIG. 5A  illustrates an error amplifier signal V EA  produced by the error amplifier  402  and a voltage sawtooth signal V sawtooth  produced by a sawtooth voltage generator (not shown) in the PWM mode of operation. When the error amplifier signal V EA  is greater then the voltage sawtooth signal V sawtooth  produced by the sawtooth voltage generator, the comparator  414  illustrated in  FIG. 4  sets the primary duty cycle D of the switching cycle. In the right portion of  FIG. 5A  is a graphical representation of the resulting inductor current I Lout  that flows through the output inductor L out . During the primary duty cycle D of the switching cycle, the inductor current I Lout  flowing through the output inductor L out  increases, and during the complementary duty cycle 1-D, the inductor current I Lout  decreases. In the right portion of  FIG. 5A , the load current (e.g., an output current I out  of the power converter) is sufficiently high so that inductor current I Lout  in the output inductor L out  does not reverse. 
     Referring now to  FIG. 5B , illustrated in the left portion of the FIGURE again is a graphical representation of the error amplifier signal V EA  produced by the error amplifier  402  and the voltage sawtooth signal V sawtooth  produced by a sawtooth voltage generator, again in the PWM mode of power converter operation. In the right portion of  FIG. 5B , the inductor current I Lout  flowing through the output inductor L out  is again illustrated. In this case, the inductor current I Lout  flowing through output inductor L out  reaches zero at the end of the switching cycle, but does not reverse direction, and thus preserving operation in the PWM mode of operation. 
     Referring now to  FIG. 5C , illustrated in the left portion of the FIGURE is a graphical representation of the error amplifier signal V EA  produced by the error amplifier  402  and the voltage sawtooth signal V sawtooth  produced by a sawtooth voltage generator in a hybrid mode of operation. In the right portion of  FIG. 5C , the inductor current I Lout  flowing through the output inductor L out  is again illustrated. In this case, the inductor current I Lout  flowing through the output inductor L out  would reach zero and reverse direction during the complementary duty cycle 1-D. In this hybrid mode of operation, the output current I out  is insufficient to prevent current reversal in output inductor L out  unless accommodation is provided in the power converter. Accommodation is provided by the AND gate  420  illustrated in  FIG. 4  that disables conduction in the auxiliary power switch Q aux  when the inductor current I Lout  flowing through the output inductor L out  reaches substantially zero or is less than zero. Thus, a controlled or remnant current level I CC  is supplied through the main power switch Q mn  to the output inductor L out , thereby maintaining the inductor current I Lout  as shown. The result is a substantial reduction of ripple current conducted to output capacitor C out  with attendant reduction in switching-induced losses in associated circuit components. 
     Referring now to  FIG. 5D , illustrated in the left portion of the FIGURE is a graphical representation of the error amplifier signal V EA  produced by the error amplifier  402  and the voltage sawtooth signal V sawtooth  produced by a sawtooth voltage generator in a light current mode of operation. In this mode, the output current I out  has been reduced even further so that output current I out  can be sustained by the controlled current level I CC  flowing through the main power switch Q mn  without alternately enabling conduction through the main and auxiliary power switches Q mn , Q aux . In this mode, a reduced voltage level for the error amplifier signal V EA  produced by the error amplifier  402  causes the error amplifier signal V EA  to lie below even the valleys of the sawtooth waveform signal V sawtooth . When the error amplifier signal V EA  lies entirely below the sawtooth waveform signal V sawtooth , the comparator  414  produces no duty cycle (e.g., disabling the duty cycle). As illustrated in the right portion of  FIG. 5D , the inductor current I Lout  that flows through the output inductor L out  is maintained at the controlled current level I CC  flowing through the main power switch Q mn , controlled by the error amplifier  402  by means of the current mirror as described previously. The result is an output current I out  controlled by the error amplifier  402  that flows to the load without active switching of either the main power switch Q mn  or the auxiliary power switch Q aux . Little to no ripple current is produced, and switching losses are substantially eliminated. Power conversion efficiency in this mode of operation is substantially improved and is determined by the ratio of output voltage V out  to input voltage V in  and remaining losses in the power converter elements. In this mode of operation, some controller elements can be selectively disabled to further reduce power losses. 
     If, in an alternative embodiment, the level of controlled current level I CC  for the main power switch Q mn  is not controlled by the error amplifier  402  in response to actual load current at light levels of load current (i.e., a preselected level of controlled current level I CC  is chosen), it is important that the preselected level of controlled current level I CC  for the main power switch Q mn  be less than the expected level of load current (again, the output current I out  of the power converter) when the load is in a low-current state. Otherwise, the output voltage V out  of the power converter can increase beyond a desired voltage level. 
     Turning now to  FIGS. 6A and 6B , illustrated are waveform diagrams of exemplary operations associated with a power switch of a power converter in accordance with the principles of the present invention. In the interest of maintaining continuity, the waveform diagrams will be described in part with reference to signals and components illustrated and described with respect to  FIG. 4 .  FIG. 6A  illustrates a graphical representation of a primary duty cycle D resulting from the error amplifier signal V EA  produced by the error amplifier  402 . When the error amplifier signal V EA  is less than a threshold level  605 , no duty cycle is produced (e.g., disabling the duty cycle). When the error amplifier signal V EA  is above the threshold level  605 , the primary duty cycle D increases linearly with the error amplifier signal V EA  until it reaches 100%. In a preferred embodiment, the error amplifier  402  is constructed to produce an error amplifier signal V EA  that can fall below the minimum voltage of the sawtooth waveform signal V sawtooth  to provide a mode of operation wherein no duty cycle is produced (e.g., disabling the duty cycle). 
     Referring now to  FIG. 6B , illustrated is a graphical representation of a controlled or remnant current level I CC  that flows through the main power switch Q mn  during, for instance, the complementary duty cycle 1-D of the switching cycle as a function of the error amplifier signal V EA  produced by the error amplifier  402 . In a preferred embodiment, above an error amplifier voltage level  610 , the controlled current level I CC  attains a saturation level  615 . In an alternative embodiment, above the error amplifier voltage level  610 , the controlled current level I CC  is reduced as the error amplifier signal V EA  increases beyond the error amplifier voltage level  610  produced by the error amplifier  402  as represented by dashed line  620 . While the controlled current level I CC  illustrated in  FIG. 6B  are represented by straight lines, it is contemplated that these lines may be implemented as nonlinear functions of the error amplifier signal V EA  to produce further efficiency enhancements associated with a particular application. 
     As mentioned above, the power converter includes a controller configured to provide a plurality of modes of operation. In a PWM mode of operation, the main and auxiliary power switches are alternately enabled to conduct in substantially complementary portions of a switching cycle in response to an output characteristic, such as an output voltage of the power converter. A beginning time of a switching cycle is ordinarily controlled by a switching cycle (e.g., frequency) clock. In the PWM mode of operation, the controller controls the level of current in the main power switch with an error amplifier with an input coupled to an output characteristic of the power converter such an output voltage. 
     The controller controls the level of current (e.g., a controlled current level such as a remnant current level) in the main power switch during the complementary duty cycle 1-D of the switching cycle when the auxiliary power switch is enabled to conduct (i.e., during an interval of time that begins substantially after the auxiliary power switch is enabled to conduct and ends substantially when the main power switch enabled to conduct). The controller may employ a current mirror to control the remnant current level in the main power switch. In the PWM mode of operation and to enhance power conversion efficiency, the controller preferably reduces the remnant current level in the main power switch during the complementary duty cycle 1-D in response to an increase of a sensed current of the power converter, such as an output current. 
     In a light current mode of operation (which may occur after the PWM mode of operation), when a current detected in the auxiliary power switch is less than a current threshold level, the controller disables the alternately enabled conduction of the main and auxiliary power switches and controls a remnant current level in the main power switch to regulate the output characteristic. In the light current mode of operation, the controller includes an error amplifier coupled to an output characteristic of the power converter such as the output voltage of the power converter to control the remnant current level in the main power switch. To enhance power conversion efficiency in the light current mode of operation, the controller selectively disables controller elements such as the operational amplifier  409  of the error amplifier  402  and the comparator  414  illustrated in  FIG. 4  and the bias sources thereto. 
     In a recharge mode of operation (which may occur following the light current mode of operation), the main and auxiliary power switches are alternately enabled to conduct in substantially complementary portions of a switching cycle in response to an output characteristic, such as an output voltage of the power converter. The controller, however, controls a conduction of the main power switch to provide a level of current (e.g., a controlled current level such as a remnant current level) in the main power switch when a level of current in the main power switch crosses an upper current threshold level, which may be a fixed upper current threshold level. The controller also enables conduction in the auxiliary power switch when the level of current in the main power switch crosses the upper current threshold level. The controller may also perform the aforementioned operations in accordance with monitoring an inductor current (as opposed or in addition to the current in the main power switch) of an output inductor of a power converter employing the main and auxiliary power switches. Thereafter, the controller disables conduction in the auxiliary power switch when a level of current in the auxiliary power switch crosses a lower current threshold level (e.g., approximately zero current) to prevent reverse current flow in an output inductor or the auxiliary power switch. In a particular application, it can be advantageous to allow a small reverse current flow in the output inductor or the auxiliary power switch to enable substantially zero voltage switching of the main power switch, thereby reducing switching losses. 
     The controller transitions between modes of operation by testing operational parameters in the power converter. In the PWM mode of operation, the controller transitions to the light current mode of operation when a level of current in the auxiliary power switch or in the output inductor is reduced to a current threshold level such as zero current when a discontinuous current mode (“DCM”) is reached. The controller transitions from the light current mode of operation to the recharge mode of operation and resets a switching cycle counter when an output characteristic falls below a lower output characteristic threshold level such as a lower output voltage threshold level. 
     In the recharge mode of operation, the controller transitions back to the PWM mode of operation when a switching cycle counter exceeds a predetermined count. In the recharge mode of operation, the controller transitions to the light current mode of operation when the output characteristic such as the output voltage exceeds an upper output characteristic threshold level, such as an upper output voltage threshold level. In the recharge mode of operation, the controller enables conduction in the main power switch with a switching cycle clock (e.g., a switching cycle clock employed to initiate a switching cycle). When the controller initially enters the recharge mode of operation, the controller initiates a count of switching cycles in accordance with a switching cycle counter. 
     Turning now to  FIG. 7 , illustrated is a flow diagram of an embodiment of a method of operating a controller in accordance with the principles of the present invention. In addition to the exemplary order that follows, it should be understood that the method may begin at one of the described modes of operation. For instance, the controller may begin in a recharge mode of operation and follow the method of operation therefrom as provided below. 
     Beginning in a PWM mode of operation, the controller samples a current, such as an auxiliary power switch current I auxsw  (or an inductor current in an output inductor of a power converter), and compares the auxiliary power switch current I auxsw  to a current threshold level I thresh , in a step or module  705 . If the auxiliary power switch current I auxsw  is less than the current threshold level I thresh  the controller sets a flag F PWM  to zero in a step or module  710  to disable the PWM mode of operation and transitions to a light current mode of operation. Otherwise, the controller continues in the PWM mode of operation and iteratively performs step  705  until the auxiliary power switch current I auxsw  exceeds the current threshold level I thresh  or the controller changes the mode of operation based on another process therein. 
     In the light current mode of operation, the controller compares an output characteristic (e.g., an output voltage V out ) of the power converter to a lower output characteristic threshold level (e.g., a lower output voltage threshold level V lower ) in a step or module  715 . If the output voltage V out  is less than the lower output voltage threshold level V lower , the controller resets a switching cycle counter, for example, to zero in a step or module  720 , and transitions to a recharge mode of operation. Otherwise, the controller continues in the light current mode of operation and iteratively performs step  715  until the output voltage V out  exceeds the lower output voltage threshold level V lower  or the controller changes the mode of operation based on another process therein 
     In the recharge mode of operation during each switching cycle, the controller increments the switching cycle counter in a step or module  725  and executes a switch logic in a step or module  730  as described below with respect to  FIG. 8 . The controller then compares the count of the switching cycle counter to a predetermined count such as a maximum count (designated “count_max”) in a step or module  735 . If the count exceeds the maximum count, the controller sets a flag F PWM  equal to one to enable the PWM mode of operation in the following switching cycle in a step or module  740  and then exits to the PWM mode of operation. Otherwise, the controller then compares an output characteristic (e.g., an output voltage V out ) of the power converter to an upper output characteristic threshold level (e.g., an upper output voltage threshold level V upper ) in a step or module  745 . If the output voltage V out  is greater than the upper output voltage threshold level V upper , the controller transitions to the light current mode of operation. Otherwise, the controller continues in the recharge mode of operation and iteratively performs steps  725 , et seq., until the controller changes the mode of operation based thereon or another process therein. 
     Turning now to  FIG. 8 , illustrated is a flow diagram of an embodiment of a method of operating a controller in accordance with the principles of the present invention. The exemplary method of  FIG. 8  demonstrates an operation of the switch logic introduced above with respect to  FIG. 7 . In a step or module  805 , the controller determines if a new switching cycle is beginning (e.g., as indicated by a time  910  illustrated in  FIGS. 9A and 9B  below). If a new switching cycle is beginning, the controller sets a flag F D  to one to enable conduction of a main power switch in a step or module  810 . If it is not the beginning of a new switching cycle, the controller determines if the flag F D  is equal to one in a step or module  815 . If the value of the flag F D  is equal to one, the controller determines if a main power switch current I auxsw  or an inductor current I Lout  in an output inductor exceeds an upper current threshold level I upper  in a step or module  820 . If the main power switch current I mnsw  or an inductor current I Lout  exceeds the upper current threshold level I upper , the flag F D  is reset to zero and a conductivity of the main power switch is reduced to provide a controlled current level (e.g., a remnant current level) in a step or module  825 . Additionally, a flag F 1-D  is set to one to enable conduction of an auxiliary power switch in the step or module  825 . If the main power switch current I mnsw  or an inductor current I Lout  does not exceed the upper current threshold level I upper , the controller continues the conduction of the main power switch and iteratively performs step  815 , et seq. 
     If the value of the flag F D  is not equal to one, the controller determines if a flag F 1-D  is equal to one in a step or module  830 . If the value of the flag F 1-D  is not equal to one, the controller returns to step  805  or performs other processes therein. If the value of the flag F 1-D  is equal to one (see also step or module  825  introduced above), the controller determines if an auxiliary power switch current I auxsw  is less than a lower current threshold level I lower  in a step or module  835 . If the auxiliary power switch current I auxsw  is less than the lower current threshold level I lower , the flag F 1-D  is reset to zero and a conductivity of the auxiliary power switch is disabled in a step or module  840 . Thereafter, the controller may return to step  805  or perform other processes therein. If the auxiliary power switch current I auxsw  is not less than the lower current threshold level I lower , the controller continues the conduction of the auxiliary power switch and iteratively performs step  830 , et seq. 
     Turning now  FIGS. 9A and 9B , illustrated are waveform diagrams of exemplary operations associated with power switches of a power converter in accordance with the principles of the present invention. The waveforms illustrate currents flowing through main and auxiliary power switches in a recharge mode of operation. The waveform  905  represents current flowing through the main power switch during a switching cycle of period ΔT s  that begins at time  910  and ends at time  915 . The current in the main power switch is enabled to conduct at the time  910  by a switching cycle clock. The current in the main power switch increases until it reaches an upper current threshold level (“I upper ”) at time  920 , at which time conductivity of the main power switch is reduced to provide a controlled current level (e.g., a remnant current level L cc ) by the controller. At the time  920 , conductivity of the auxiliary power switch is enabled to conduct as represented by the waveform  925 . The current in the auxiliary power switch decreases until it crosses a lower current threshold level (“I lower ”) at time  930 , and remains substantially zero until the end of the switching cycle at time  915 . 
     Turning now to  FIG. 10 , illustrated is a flow diagram of an embodiment of a method of operating a controller in accordance with the principles of the present invention. In addition to the exemplary order that follows, it should be understood that the method may begin at one of the described modes of operation. For instance, the controller may begin in a recharge mode of operation and follow the method of operation therefrom as provided below. 
     Beginning in a PWM mode of operation, the controller samples a current, such as an auxiliary power switch current I auxsw  (or an inductor current in an output inductor of a power converter), and compares the auxiliary power switch current I auxsw  to a current threshold level I thresh , in a step or module  1005 . If the auxiliary power switch current I auxsw  is less than the current threshold level I thresh , the controller sets a flag F PWM  to zero in a step or module  1010  to disable the PWM mode of operation and transitions to a light current mode of operation. Otherwise, the controller continues in the PWM mode of operation and iteratively performs step  1005  until the auxiliary power switch current I auxsw  exceeds the current threshold level I thresh  or the controller changes the mode of operation based on another process therein. 
     Accordingly, the controller can advantageously transition to the light current mode of operation when DCM is reached. A controlled current level (e.g., a remnant current level) in the main power switch that is typically a small level of current is controlled in an interval of time that begins substantially when the auxiliary power switch is enabled to conduct and ends substantially when the main power switch is enabled to conduct (i.e., during the complementary duty cycle 1-D). The controller controls the remnant current level employing a current mirror in response to an output characteristic of the power converter. The controller may reduce the remnant current level in response to an increase of a sensed current of the power converter, such as a sensed output current. 
     In the light current mode of operation, the controller compares an output characteristic (e.g., an output voltage V out ) of the power converter to a lower output characteristic threshold level (e.g., a lower output voltage threshold level V lower ) in a step or module  1015 . If the output voltage V out  is less than the lower output voltage threshold level V lower , the controller resets a switching cycle counter, for example, to zero in a step or module  1020 , and transitions to a recharge mode of operation. Otherwise, the controller continues in the light current mode of operation and iteratively performs step  1015  until the output voltage V out  exceeds the lower output voltage threshold level V lower  or the controller changes the mode of operation based on another process therein. 
     In the recharge mode of operation, the main and auxiliary power switches are alternately enabled to conduct, and conduction in the main power switch is reduced to a remnant current level when a level of current in the main power switch (or an output inductor) increases to an upper current threshold level, which may be a fixed or adjustable upper current threshold level. Conduction in the auxiliary power switch is terminated when a level of current in the auxiliary power switch is reduced to a lower current threshold level, which may be a fixed or adjustable lower current threshold level of substantially zero current. The remnant current level is controlled by the controller responsive to the output characteristic of the power converter. In the recharge mode of operation, the controller initiates conduction by the main power switch with the switching cycle clock. 
     In the recharge mode of operation during each switching cycle, the controller compares the count of a switching cycle counter to a predetermined count such as a maximum count (designated “count_max”) in a step or module  1025 . If the count exceeds the maximum count, the controller transitions to a hybrid mode of operation. Otherwise, the controller executes a switch logic in a step or module  1030  as described above with respect to  FIG. 8 . The controller then compares an output characteristic (e.g., an output voltage V out ) of the power converter to an upper output characteristic threshold level (e.g., an upper output voltage threshold level V upper ) in a step or module  1035 . If the output voltage V out  is greater than the upper output voltage threshold level V upper , the controller transitions to the light current mode of operation. Otherwise, the controller increments the switching cycle counter in a step or module  1040 , continues in the recharge mode of operation and iteratively performs steps  1025 , et seq., until the controller changes the mode of operation based thereon or another process therein. 
     In the hybrid mode of operation, the main and auxiliary power switches are alternately enabled to conduct in the substantially complementary portions of the switching cycle. The alternately enabled conduction is ordinarily controlled by an error amplifier coupled to the output characteristic of the power converter. In the hybrid mode of operation, conduction in the auxiliary power switch is terminated when a level of current in the auxiliary power switch is reduced to a lower current threshold level. Conduction in the auxiliary power switch is terminated to prevent a reverse current flowing through the output inductor of the power converter. A small reverse current flowing through the output inductor may be allowed to enable soft switching of the main power switch. A remnant current level in the main power switch is controlled in an interval of time that begins substantially when the auxiliary power switch is enabled to conduct and ends substantially when the main power switch is enabled to conduct (i.e., during a complementary duty cycle 1-D). The controller employs an error amplifier coupled to the output terminal of the power converter to control the remnant current level. In an exemplary case, when an output characteristic crosses a lower output characteristic threshold in the hybrid mode of operation, the switching cycle is terminated prior to the start time of the next switching cycle that is ordinarily controlled by the switching cycle clock. In another exemplary case, when a level of the output characteristic increases to an upper output characteristic threshold in the hybrid mode of operation, conduction in the main power switch can also be reduced to the remnant current level prior to a main power switch conduction termination time controlled by the error amplifier (i.e., during the primary duty cycle D). 
     In the hybrid mode of operation, the controller compares an output characteristic (e.g., an output voltage V out ) of the power converter to an upper output characteristic threshold level (e.g., an upper output voltage threshold level V upper ) in a step or module  1045 . If the output voltage V out  is greater than the upper output voltage threshold level V upper , the flag F D  is reset to zero and a conductivity of the main power switch is reduced to provide a controlled current level (e.g., a remnant current level) in a step or module  1050 . Additionally, a flag F 1-D  is set to one to enable conduction of an auxiliary power switch in the step or module  1050 . Thereafter, and if the output voltage V out  is not greater than the upper output voltage threshold level V upper , the controller compares an output characteristic (e.g., the output voltage V out ) of the power converter to a lower output characteristic threshold level (e.g., a lower output voltage threshold level V lower ) in a step or module  1055 . If the output voltage V out  is less than the lower output voltage threshold level V lower , the controller terminates the switching cycle prior to the start time of the next switching cycle in a step or module  1060 . Thereafter, or if the output voltage V out  is not less than the lower output voltage threshold level V lower , the controller compares a time (e.g., a duration of time operating in the hybrid mode of operation) to a predetermined or a maximum time (designated “timeout_max”) in a step or module  1065 . The maximum time may be computed by adding a time increment, such as 600 microseconds, to the time when the controller enters the hybrid mode of operation. If the time is greater than the maximum time, the controller sets a flag F PWM  equal to one in a step or module  1070  to signal a transition to the PWM mode of operation in the next switching cycle. Otherwise, the controller continues in the hybrid mode of operation and iteratively performs step  1045 , et seq. for the time window defined by the maximum time. 
     In an exemplary embodiment, adaptation of a duty cycle is made from time to time in, for instance, a PWM mode of operation to enable detection of a reverse or low level of current flow in an auxiliary power switch or output inductor. In conventional PWM operation, when the output voltage of the power converter is not significantly lower than the input voltage (which in a buck power converter generally produces a very short duty cycle for the auxiliary power switch), there may be insufficient time for current to flow in the auxiliary power switch for the controller to make an accurate determination of auxiliary power switch current. Accordingly, the controller may not make an accurate or reliable determination of the need to enter into the light current mode of operation. To provide a sufficient interval of time to enable the controller to measure the auxiliary power switch current, the controller from time to time limits a duty cycle of the auxiliary power switch, preferably for one switching cycle. For example, the controller may limit a duty cycle for one switching cycle for the auxiliary power switch periodically every sixty-four switching cycles and a corresponding increase a duty cycle for the main power switch during that switching cycle to preserve a switching frequency. The duty cycle of the auxiliary power switch is accordingly periodically limited, preferably for one switching cycle, after a number of switching cycles. Such infrequent modification of a duty cycle, which need not be periodic, produces a small deviation in an output characteristic such as an output voltage from a desired regulated value, which is generally not detrimental to the operation of the power converter. 
     Turning now to  FIG. 11 , illustrated is a flow diagram of an embodiment of a method of operating a controller in accordance with the principles of the present invention. In addition to the exemplary order that follows, it should be understood that the method may begin at one of the described modes of operation. For instance, the controller may begin in a recharge mode of operation and follow the method of operation therefrom as provided below. 
     In the PWM mode of operation, the main and auxiliary power switches are alternately enabled to conduct in substantially complementary portions of a switching cycle in response to an output characteristic such as an output voltage of the power converter. A duty cycle is periodically limited for one switching cycle to facilitate detection of a reverse or low level of current flow in an auxiliary power switch or in an output inductor. If a switching cycle counter is greater than a predetermined count such as a maximum count (designated “count_max”) as illustrated in a step or module  1105 , then the duty cycle of a main power switch is increased (e.g., to a maximum value) and, correspondingly, the duty cycle of the auxiliary power switch is limited (e.g., to a minimum value) in a step or module  1110 . For instance, the predetermined or maximum count may correspond to 64 switching cycles. 
     Thereafter, the controller samples a current, such as an auxiliary power switch current I auxsw  (or an inductor current in an output inductor of a power converter), and compares the auxiliary power switch current I auxsw  to a current threshold level I thresh  in a step or module  1115 . If the auxiliary power switch current I auxsw  is less than the current threshold level I thresh , the controller sets a flag F PWM  to zero in a step or module  1120  to disable the PWM mode of operation and transitions to a light current mode of operation. Otherwise, or if the switching cycle counter is not greater than a maximum count in accordance with step or module  1105 , the switching cycle counter (“count”) is incremented as illustrated in a step or module  1125  and the controller continues in the PWM mode of operation and iteratively performs steps  1105 , et seq. until the controller changes the mode of operation based thereon or another process therein. In this manner, a duty cycle is periodically limited for one switching cycle. Of course, the duty cycle limit need not be periodic, and the limit may be applied for more than one switching cycle to facilitate the current detection. 
     In the light current mode of operation, the controller compares an output characteristic (e.g., an output voltage V out ) of the power converter to a lower output characteristic threshold level (e.g., a lower output voltage threshold level V lower ) in a step or module  1130 . If the output voltage V out  is less than the lower output voltage threshold level V lower , the controller resets a switching cycle counter, for example, to zero in a step or module  1140 , and transitions to a recharge mode of operation. Otherwise, the controller continues in the light current mode of operation and iteratively performs step  1130  until the output voltage V out  exceeds the lower output voltage threshold level V lower  or the controller changes the mode of operation based on another process therein. 
     In the recharge mode of operation during each switching cycle, the controller compares the count of a switching cycle counter to a predetermined count such as a maximum count (designated “count_max”) in a step or module  1150 . If the count exceeds the maximum count, the controller sets a flag F PWM  equal to one in a step or module  1155 , resets a switching cycle counter, for example, to zero in a step or module  1160 , and transitions to the PWM mode of operation. It should be noted that the maximum count that enables the transition from the recharge mode of operation to the PWM mode of operation may be different from the maximum count employed in the PWM mode of operation to limit a duty cycle of a power switch. 
     Otherwise, the controller executes a switch logic in a step or module  1165  as described above with respect to  FIG. 8 . The controller then compares an output characteristic (e.g., an output voltage V out ) of the power converter to an upper output characteristic threshold level (e.g., an upper output voltage threshold level V upper ) in a step or module  1170 . If the output voltage V out  is greater than the upper output voltage threshold level V upper , the controller transitions to the light current mode of operation. Otherwise, the controller increments the switching cycle counter in a step or module  1175 , continues in the recharge mode of operation and iteratively performs steps  1150 , et seq., until the controller changes the mode of operation based thereon or another process therein. 
     Turning now to  FIG. 12 , illustrated is a schematic drawing of an embodiment of portions of a power converter constructed according to the principles of the present invention. The power converter includes a controller having an error amplifier  1205  that compares a desired system voltage V system  with an output voltage V out  of the power converter. A more complete schematic drawing of an error amplifier was illustrated and described above with respect to  FIG. 4 . An error amplifier signal  1210  from the error amplifier  1205  is coupled to control subsystem  1215  that includes mode transition logic. A first comparator  1220  compares the output voltage V out  of the power converter with an upper output voltage threshold level V upper  to produce a first comparator signal  1225 , which is coupled to the control subsystem  1210  to enable a mode transition from a recharge mode of operation to a light current mode of operation. 
     A second comparator  1230  compares the output voltage V out  of the power converter with a lower output voltage threshold level V lower  to produce a second comparator signal  1235 , which is coupled to the controller subsystem  1215  to enable a mode transition from a light current mode of operation to a recharge mode of operation. A third comparator  1240  compares a sensed inductor current I Lout  in an output inductor L out  with a current threshold level I thresh  to produce a third comparator signal  1245 , which is coupled to the controller subsystem  1215  to enable a mode transition from a PWM mode of operation to a light current mode of operation. In another embodiment, the third comparator  1240  may compare a sensed current in a power switch with the current threshold level I thresh  to produce the third comparator signal  1245 . In an analog circuit implementation, the threshold levels may be produced with a voltage reference and voltage dividers. In a digital circuit implementation, the threshold voltages may be recorded as stored data for use by digital logic to make the comparisons. 
     The controller also illustrates a summer  1250  that subtracts a signal at a circuit node  1255  representing a current level from the current threshold level I thresh . As illustrated in  FIG. 12 , this signal representing the current level is produced by a resistor divider network formed with resistors  1260 ,  1265  that proportionately divide a voltage difference produced by a positive bias voltage source Vdd and a negative bias voltage source −Vdd. Accordingly, the signal at the circuit node  1255  representing the current level can be produced with positive or negative values according to the ratio of the resistance of the resistors  1260 ,  1265 . The effect of subtraction of this signal at the circuit node  1255  representing the current level from the current threshold level I thresh  enables adjustment of a current level at which conductivity of an auxiliary power switch Q aux  is disabled (i.e., field adjustable to meet an application). The resistor  1260  may be provided as an external resistor coupled to external terminals to enable end users to perform this adjustment on their circuit boards without the need to modify the controller. Inclusion of an external resistor to perform this adjustment may be employed, without limitation, with any of the embodiments described hereinabove to control a mode transition from, for instance, a PWM mode of operation to a light current mode of operation. In accordance with the foregoing, the controller provides gate drive signals S DRV1 , S DRV2  to control conductivity of the main and auxiliary power switches Q mn , Q aux , respectively, of the power converter. 
     Turning now to  FIG. 13 , illustrated is a schematic drawing of an embodiment of portions of a power converter constructed according to the principles of the present invention. The power converter includes a controller that may periodically limit a duty cycle of a power switch in the power converter to facilitate detecting an auxiliary power switch current in an auxiliary power switch Q aux . A control subsystem  1310  including mode transition logic produces initial gate drive signals S DRV1 , S DRV2  to control conductivity of the main and auxiliary power switches Q mn , Q aux  respectively, of the power converter. The controller also includes a duty cycle subsystem  1320  that produces gate drive signals S DRV1 , S DRV2  from the initial gate drive signals S DRV1 , S DRV2  and a switching cycle clock signal (e.g., divide by 64) to periodically set or limit a duty cycle of at least one of the main and auxiliary power switches Q mn , Q aux . The duty cycle subsystem  1320  may be formed from comparator logic akin to the controller illustrated and described with respect to  FIG. 12 . In accordance with the switching cycle clock signal, the duty cycle subsystem  1320  counts a number of switching cycles after which it limits a duty cycle of a power switch such as a one-cycle power switch duty cycle limit. 
     Turning now to  FIG. 14 , illustrated is a schematic diagram of an embodiment of portions of a power converter constructed according to the principles of the present invention. The power converter includes a controller  1400  that regulates an output characteristic (e.g., an output voltage V out ) of the power converter, and is configured to control current through a main power switch Q mn  in response to a sensed or estimated current of the power converter. The controller  1400  of the power converter is operable in a plurality of modes of operation as described above. The logical processes to perform the modes of operation and transitions therebetween may be formed with digital circuit elements in accordance with a switching cycle clock to set flags to control circuit elements illustrated in  FIG. 14 . The controller  1400  is formed with elements illustrated and described hereinabove with reference to  FIG. 4  that will not be redescribed in the interest of brevity. 
     Again, in addition to the elements described with respect to  FIG. 4 , the controller  1400  includes an AND gate  1405  that produces a signal to enable conductivity of main and auxiliary power switches Q mn , Q aux . If a flag F PWM  is set to one (meaning that the controller is in a PWM mode of operation), then the AND gate  1405  produces a signal that is coupled to the input of an OR gate  1410  that enables a pulse-width modulated signal S PWM  to control alternating conductivity of the main and auxiliary power switches Q mn , Q aux  in response to an error amplifier signal V EA  produced by an operational amplifier  409 . If the flag F PWM  is set to zero, for example, in a recharge mode of operation, then an inductor current I Lout  sensed by a current sensor  460  exceeding an upper current threshold level I upper  is detected by a comparator  1415 , and the resulting signal is coupled through an OR gate  1420  to an input of the AND gate  1405  to disable the pulse-width modulated signal S PWM . Accordingly, conductivity of the main power switch Q mn  is reduced to a remnant current level as described above. Continuing in the recharge mode of operation, when the output voltage V out  is lower than the upper output voltage threshold level V upper  and the flag F PWM  is set to zero, a flag F D  produced by a switch logic (see, e.g., switch logic illustrated and described with respect to  FIG. 8 ) controls the alternating conductivity of the main and auxiliary power switches Q mn , Q aux  in conjunction with a comparator  1425 . 
     Thus, the controller is constructed to provide a PWM mode of operation for a power converter with the main and auxiliary power switches Q mn , Q aux  alternately enabled to conduct in substantially complementary portions of a switching cycle in response to an output characteristic of the power converter such as the output voltage V out . The controller also provides a recharge mode of operation for the power converter wherein the main and auxiliary power switches Q mn , Q aux  are alternately enabled to conduct, and conduction in the main power switch Q mn  is terminated when a level of current, for example, a main power switch current in the main power switch Q mn  or the inductor current I Lout  in the output inductor L out  increases to the upper current threshold level I upper . The current sensor  460  mat sense a level of current in the output inductor L out , the main power switch Q mn  or the auxiliary power switch Q aux  depending on which switch is enabled to conduct. Conduction in the auxiliary power switch Q aux  is terminated when an auxiliary power switch current in the auxiliary power switch Q aux  is reduced to a lower current threshold level. The controller controls a remnant current level in the main power switch Q mn  during the complementary duty cycle  1 -D. 
     The controller also provides a hybrid mode of operation wherein the main and auxiliary power switches Q mn , Q aux  are alternately enabled to conduct in the substantially complementary portions of the switching cycle controlled by an error amplifier coupled to the output characteristic of the power converter. Conduction in the main power switch Q mn  is reduced to a remnant current level prior to a main power switch conduction termination time controlled by the error amplifier when a level of the output characteristic (e.g., the output voltage V out ) increases to an upper output characteristic threshold level (e.g., the upper output voltage threshold level V upper ) in conjunction with the comparator. Conduction in the auxiliary power switch Q aux  is terminated prior to the control by the error amplifier when the auxiliary power switch current in the auxiliary power switch Q aux  is reduced to a lower current threshold level. The switching cycle is terminated prior to the end time controlled by the switching cycle clock when the output characteristic such as the output voltage V out  crosses a lower output characteristic threshold level, and a remnant current level in the main power switch Q mn  is controlled in an interval of time that begins substantially when the auxiliary power switch Q aux  is enabled to conduct, and ends substantially when the main power switch Q mn  is enabled to conduct (i.e., during the complementary duty cycle). A duty cycle of the auxiliary power switch Q aux  is periodically limited for at least one of the switching cycles after a number of switching cycles. 
     Turning now to  FIG. 15 , illustrated is a schematic drawing of an embodiment of portions of a controller constructed according to the principles of the present invention. The controller includes comparator  1510  (see, e.g., comparator  414  illustrated and described with respect to  FIG. 4 ). A switch formed as a P-channel MOSFET Q15 is coupled in series with a positive bias node of the comparator  1510  and the positive bias voltage source Vdd. When a flag F PWM  exhibits a positive voltage (i.e., when the value of the flag F PWM  is one to indicate that the power converter is operating in a PWM mode of operation), an inverter  1520  applies a substantially zero voltage to the gate of the P-channel MOSFET Q15, thereby turning on the same. When the flag F PWM  exhibits substantially zero voltage (i.e., when the value of the flag F PWM  is zero), the inverse situation occurs (i.e., the positive bias voltage source Vdd is decoupled from the comparator  1510 ), thereby disabling operation of the comparator  1510 . The flag F PWM  is equal to zero in the light current mode of operation. In this manner, switching and other inherent losses in the comparator  1510  can be avoided when the power converter is operating in the light current mode of operation. Of course, other logic may be included in the design of the controller to enable or disable operation of the comparator  1510  in another mode of operation such as a recharge mode of operation. 
     In applications of a power converter for which an end user desires to improve power converter efficiency when the power converter delivers a light current to a load, it is advantageous to adjust a current threshold level at which the power converter transitions from a PWM mode of operation to a light current mode of operation. In the light current mode of operation, the alternately enabled conductivity of the main and auxiliary power switches is disabled when current flowing in a power switch such as the auxiliary power switch or in the output inductor falls below the current threshold level. In the PWM mode of operation, the magnitude of the peak-to-peak ripple current I pk     —     pk  produced in the output inductor is dependent on the input voltage V in  and the output voltage V out  of the power converter. Accordingly, the input voltage V in  and the output voltage V out  determine a direct current level of an output current I outdc  from the power converter at which the power converter transitions from the PWM mode of operation to the light current mode of operation. 
     The value of the magnitude of the peak-to-peak ripple current I pk     —     pk  can be calculated for a buck power converter according to equation (3): 
                       I   pk_pk     =             V   in     -     V   out         L   out       ·     t   ON       =         V   out       L   out       ·     t   OFF           ,           (   3   )               
where
 
     t ON  is the on-time of the main power switch (i.e., t ON =D·ΔT s ), 
     t OFF  is the complement of the on-time of the main power switch with respect to the period ΔT s  of a switching cycle (i.e., t OFF =(1-D)·ΔT s ) and 
     L out  in equation (1) is the inductance of the output inductor. 
     The parameter ΔT s  is the period of a switching cycle as illustrated in  FIGS. 9A and 9B . During the on-time t ON  of the main power switch, the current in the output inductor has a positive slope, and during the on time of the auxiliary power switch, the current in the output inductor has a negative slope. Accordingly, at the end of the on time of the auxiliary power switch, the auxiliary power switch current and the inductor current in the output inductor reaches a minimum value. 
     The minimum value I minLout  of the instantaneous current in the output inductor or the auxiliary power switch can be calculated from the direct current level of an output current I outdc  and the on-time t ON  according to equation (4): 
                     I     min   ⁢           ⁢   Lout       =       I   outdc     -           V   in     -     V   out         2   ·     L   out         ·       t   ON     .                 (   4   )               
If the end user desires the power converter to transition from the PWM mode of operation to the light current mode of operation at a particular direct current level of an output current I outdc  of the power converter, then the current threshold level I thresh  illustrated in  FIG. 12  is adjusted (with an offset as desired produced by the signal at the circuit node  1255 ) to the minimum value I minLout  corresponding to the direct current level of an output current I outdc  and the on-time t ON  as indicated above by equation (2).
 
     Turning now to  FIG. 16 , illustrated is a schematic drawing of an embodiment of portions of a controller constructed according to the principles of the present invention. The controller is formed with some of the elements illustrated and described hereinabove with reference to  FIG. 12  that will not be redescribed in the interest of brevity. The controller is configured to adjust a current threshold level I thresh  in accordance with the direct current level of an output current from the power converter at which the controller transitions from the PWM mode of operation to the light current mode of operation. A threshold subsystem  1605  produces a current threshold level I thresh  that the controller employs to set a flag F PWM  to zero (see, e.g., step or module  710  in  FIG. 7 ) to disable a PWM mode of operation and to transition to the light current mode of operation. The threshold subsystem  1605  generates or computes the current threshold level I thresh  according to equation (4) from a signal at a circuit node  1610  that represents the direct current level an output current from the power converter at which the controller transitions from the PWM mode of operation to the light current mode of operation. 
     The controller also employs a signal  1615  that represents the duty cycle D of the main power switch Q mn  that determines the on-time t ON . The controller internally computes the duty cycle D in determination of the pulse-width modulated signal S PWM  illustrated in  FIGS. 4 and 14  and, can thus, readily access the on-time t ON . The direct current level of an output current is set by the resistor divider network formed with resistors  1620 ,  1630  coupled between the positive bias voltage source Vdd and circuit ground. The resistor  1630  may be an external resistor coupled between external nodes to enable an end user to set (or adjust) the desired direct current level of the output current. Thus, the direct current level of the output current and the on-time input are provided to the threshold subsystem  1605  to enable computation of the current threshold level I thresh  to control the direct current level of the output current at which the controller transitions from the PWM mode of operation to the light current mode of operation. 
     Thus, as illustrated and described with reference to the accompanying drawings, a controller for a circuit such as a switch-mode power converter operable in a plurality of modes is constructed in an advantageous embodiment. The controller may advantageously be formed with an error amplifier that is coupled to power converter elements to disable power switch conductivity upon, for instance, reversal or reduction of a power converter current, and to provide a controlled current level in a main power switch. It is contemplated within the broad scope of the invention that a mode of operation can be changed based upon a power converter current, sensed or estimated, falling below a threshold level. It is further contemplated that a controller can be constructed with a plurality of error amplifiers to control an output characteristic in a plurality of modes of operation. It is further contemplated that the controller can be constructed to operate in a plurality of modes of operation based on sensing or estimating a parameter that may be an indirect indicator of an output characteristic such as an output current. 
     Those skilled in the art should understand that the previously described embodiments of a power converter and related methods of constructing the same are submitted for illustrative purposes only. In addition, other embodiments capable of producing a power converter employable with other switch-mode power converter topologies are well within the broad scope of the present invention. While the power converter has been described in the environment of a power converter including a controller to control an output characteristic to power a load, the power converter including a controller may also be applied to other systems such as a power amplifier, a motor controller, and a system to control an actuator in accordance with a stepper motor or other electromechanical device. 
     For a better understanding of power converters, see “Modern DC-to-DC Switchmode Power Converter Circuits,” by Rudolph P. Severns and Gordon Bloom, Van Nostrand Reinhold Company, New York, N.Y. (1985) and “Principles of Power Electronics,” by J. G. Kassakian, M. F. Schlecht and G. C. Verghese, Addison-Wesley (1991). The aforementioned references are incorporated herein by reference in their entirety. 
     Also, although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. For example, many of the processes discussed above can be implemented in different methodologies and replaced by other processes, or a combination thereof. 
     Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, composition of matter, means, methods, and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.