Patent Publication Number: US-7218692-B2

Title: Multi-path interference cancellation for transmit diversity

Description:
This application claims the priority under 35 U.S.C. 119(e)(1) of copending U.S. provisional application No. 60/298,784, filed on Jun. 15, 2001, and incorporated herein by reference. 

   BACKGROUND OF THE INVENTION 
   1. Technical Field of the Invention 
   The present invention relates to wireless telecommunications and, more particularly, to multi-path interference cancellation in a high speed data system using transmit diversity. 
   2. Description of Related Art 
   A phenomena that reduces the efficiency of a communications link is fading. Fading may take several forms, one of which is referred to as multi-path fading. Multi-path fading is caused by two or more copies of a transmitted signal combining at the receiver in a way that reduces the overall received signal level. One technique developed for reducing the effects of fading is transmit diversity. Generally, for transmit diversity, a radio transmitter uses two transmit antennas that are positioned far from each other to transmit one signal. Typically, the two transmit antennas are positioned more than several wavelengths of the transmitted signal from each other depending upon the environment. 
   For example, in the well known Space-Time Transmit Diversity (STTD) system, symbols S 1  and S 2  are received for transmission at the transmitter encoder in which S 1  is received during the period from T 0  to T 1  and S 2  is received during the period from T 1  to T 2 . At a first output of the encoder, S 1  is output for transmission to a first antenna during the symbol time from T 1  to T 1 , followed by symbol S 2  from symbol time T 1  to T 2 . A second output of the encoder outputs the negative complex conjugate of symbol S 2  for transmission to a second antenna during time T 0  to T 1 , followed by the complex conjugate of symbol S 1  from the period T 1  to T 2 . 
   Another improvement, known as High Speed Downlink Packet Access (HSDPA), has been developed to enhance mobile services for high-speed data users. HSDPA takes advantage of link adaptation such as adaptive modulation and coding to enhance data rates to data users in a time-multiplexed manner. HSDPA is specified by in Third Generation Partnership Project (3GPP); Technical Specification Group Radio Access Network; Physical layer aspects of UTRA High Speed Downlink Packet Access (Release 4), the description of which is hereby incorporated by reference. The outcome of the 3GPP work is a set of specifications defining the 3G-network functionality, procedures and service aspects. HSDPA transmissions are performed on physical channels shared by other users generally employing different spreading gains. The HSDPA channels employ a spreading gain of 16 and voice users, for example, typically employ a spreading gain of 64. 
   HSDPA can be transmitted in a transmit diversity manner with other user transmissions. However, some conventional signal processing methods cannot be used for receivers of HSDPA data encoded in a transmit diversity scheme and, thus innovative processing methods must be developed. 
   For example, HSDPA can be encoded for STTD transmission on a shared channel with voice users in which the voice users may or may not be STTD encoded. In either case, however, because the spreading gain of the voice users is different from the HSDPA channels, the effective channel seen by the voice users is different from the channel seen by the HSDPA users. If STTD encoded according to Third Generation Partnership Project (3GPP); Technical Specification Group Radio Access Network; Physical channels and mapping of transport channels onto physical channels (FDD) (Release 1999), the description of which is hereby incorporated by reference, the STTD encoded data itself does not see a time invariant channel and, thus chip level equalization cannot be employed to remove the multi-path interference. 
   SUMMARY OF THE INVENTION 
   The present invention achieves technical advantages as an apparatus, system and method for multi-path interference cancellation for high speed data signals encoded for transmit diversity. Interference cancellation is implemented using the spreading gain of the high speed data signal, such as the spreading gain of 16 typically used for High Speed Downlink Packet Access (HSDPA). Even though the spreading gain of other user channels are 64 or greater, the root code of length  16  from these codes can be employed for linear interference cancellation. Hence, interference cancellation is implemented with a spreading gain of HSDPA, which is only length  16 . Alternatively, despreading of length 64 is implemented for the voice users when the mobile receiver has the requisite knowledge of voice users transmitting. For systems with  2  receive antennas, MTA-MPIC can also differentiate between the other users being transmitted on the multiple transmit antennas. Thus, for  2  receive antennas, MTA-MPIC can also perform a MIMO type interference cancellation for the other users. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention, reference is made to the following detailed description taken in conjunction with the accompanying drawings wherein: 
       FIG. 1A  illustrates a HSDPA/transmit diversity system; 
       FIG. 1B  illustrates STTD encoding according to release &#39;99 for HSDPA channels; 
       FIG. 1C  illustrates STTD encoding combined with orthogonal code spreading. 
       FIG. 2  shows a simple block diagram illustrating MTA-MPIC with one receive antenna and a HSDPA data user; 
       FIG. 3  shows a more detailed block diagram illustrating MTA-MPIC for the case of one receive antenna in which other user interference is not canceled; 
       FIG. 4  illustrates an alternative embodiment for MTA-MPIC with one receive antenna and a HSDPA data user in which other user interference can be canceled on a per “finger” basis; 
       FIG. 5  illustrates a block diagram of MTA-MPIC with two receive antennas and a HSDPA data user; 
       FIG. 6  illustrates an alternative embodiment for MTA-MPIC with two receive antennas and a HSDPA data user; and 
       FIG. 7  illustrates a MIMO decoder operation. 
       FIG. 8  diagrammatically illustrates pertinent portions of further exemplary embodiments of the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The numerous innovative teachings of the present application will be described with particular reference to the presently preferred exemplary embodiments. However, it should be understood that this class of embodiments provides only a few examples of the many advantageous uses and innovative teachings herein. In general, statements made in the specification of the present application do not necessarily delimit any of the various claimed inventions. Moreover, some statements may apply to some inventive features, but not to others. 
   Throughout the drawings, it is noted that the same reference numerals or letters will be used to designate like or equivalent elements having the same function. Detailed descriptions of known functions and constructions unnecessarily obscuring the subject matter of the present invention have been omitted for clarity. 
     FIG. 1B  is a simple illustration of STTD encoding according to release &#39;99 for HSDPA channels. Symbols S 1  and S 2  are received for transmission at the transmitter encoder  11  in which S 1  is received during the period from T 0  to T 1  and S 2  is received during the period from T 1  to T 2 . At a first output  13  of the encoder, S 1  is output for transmission to a first antenna during the symbol time from T 1  to T 1 , followed by symbol S 2  from symbol time T 1  to T 2 . A second output  15  of the encoder outputs the negative complex conjugate of symbol S 2  for transmission to a second antenna during time T 1  to T 1 , followed by the complex conjugate of symbol S 1  from the period T 1  to T 2 . 
   Referring now to  FIG. 1A  there is illustrated a HSDPA/transmit diversity system in which desired HSDPA channels and other users, such as control and voice users, are code multiplexed for transmission and the HSDPA channels are transmit diversity encoded. Data stream X 1 (n) comprises the HSDPA channels for antenna  1  with a spreading gain of 16 and voice and other users V 1 (n) which may have a spreading gain larger than 16. Since the HSDPA channels are transmit diversity encoded, the HSDPA data stream on antenna  2  is written by X 2 (n). The other user channels which may or may not be transmit diversity encoded are given by the data stream V 1 (n) on antenna  1 . The data stream X 1 (n) is given by the following; 
   
     
       
         
           
             
               
                 
                   
                     X 
                     1 
                   
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       m 
                       = 
                       1 
                     
                     M 
                   
                   ⁢ 
                   
                     
                       
                         C 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     ⁢ 
                     
                       
                         S 
                         1 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 1 
               
             
           
         
       
     
   
   Where C m (n) is the spreading code chip for the m th  user at time n including the long code and the Walsh code and S 1   m (n) is the symbol value of the m th  user at time n on antenna  1 . 
   The data stream for other users on antenna  2 , V 2 (n), is related to the data stream V 1 (n) on antenna  1 . The relation depends upon the spreading gain of the other users which is typically greater than a spreading gain of 16. For example, the spreading gain is typically 64 for voice users but it can be different for other applications. The relationship of V 2 (n) with V 1 (n) further depends upon the type of transmit diversity encoding used, whether open loop encoding (i.e., STTD encoding) or closed loop encoding is being employed on V 1 (n)). Hence, in general, the HSDPA is not able to find out the exact relationship of V 2 (n) with respect to V 1 (n). Therefore, the other user data stream V 2 (n) on antenna  2  should be considered independent from the data stream V 1 (n) on antenna  1  at the mobile receiver. The data stream X 2 (n), which comprises the HSDPA channels and the voice and other users V 2 (n), is given by the following; 
   
     
       
         
           
             
               
                 
                   
                     X 
                     2 
                   
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       m 
                       = 
                       1 
                     
                     M 
                   
                   ⁢ 
                   
                     
                       
                         C 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     ⁢ 
                     
                       
                         S 
                         2 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 2 
               
             
           
         
       
     
   
   Where C m (n) is the spreading code chip for the m th  user at time n including the long code and the Walsh code and S 2   m (n) is the symbol value of the m th  user at time n on antenna  2 . The spreading code employed by both the antennas is the same hence it is denoted by the same spreading sequence C m (n). 
   As above-mentioned, HSDPA channels are transmit diversity encoded and employ a spreading gain of 16 whereas voice users typically employ a spreading gain of 64 and may or may not be transmit diversity encoded. In either case, because the spreading gain of the voice users is different from the HSDPA channels, the effective channel seen by the voice users is different from the channel seen by the HSDPA users. More particularly, if STTD encoded according to release 99, the STTD encoded data itself does not see a time invariant channel. This means conventional chip level equalization cannot be employed in a mobile receiver to remove multi-path interference at the receiver. 
     FIG. 1C  illustrates a conventional example of the use of STTD in combination with orthogonal spreading codes (corresponding generally to C m (n) above) such as used in conventional CDMA systems. 
   The present invention uses multi-path interference cancellation, instead of equalization techniques, to remove the interference due to multi-path for multiple transmit antennas (hereinafter referred to as MTA-MPIC). An advantage of MTA-MPIC is that even though the spreading gain of voice users is 64, a linear decision can be made on them using a simple spreading of only length  16 . For spreading gains of other channels at 64 or greater, the root code of length  16  from these codes is used for linear interference cancellation. Hence, it is possible to implement the interference cancellation with a spreading gain of HSDPA, which is only length  16 . Alternatively, despreading of length 64 is implemented for the voice users for example, when the mobile receiver has the knowledge of which voice users are transmitting. In equations 1 and 2 without loss of generality let m=1 to m={tilde over (M)}&lt;M be the codes allocated to the HSDPA users. Then the HSDPA signals S 1   m (n), S 2   m (n); m&lt;{tilde over (M)} can be space time (ST) encoded as shown in  FIGS. 1B and 1C . 
   Letting T c  be the chip period, the spreading gain for the HSDPA users is given by K=T/T c . Letting α j   1  and α j   2  be the fading parameters from the two antennas respectively for the j th  path the net received signal at the mobile after the receive matched filter is given by; 
   
     
       
         
           
             
               
                 
                   r 
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       j 
                       = 
                       1 
                     
                     L 
                   
                   ⁢ 
                   
                     ( 
                     
                       
                         
                           α 
                           j 
                           1 
                         
                         ⁢ 
                         
                           
                             X 
                             1 
                           
                           ⁡ 
                           
                             ( 
                             
                               n 
                               - 
                               
                                 
                                   τ 
                                   j 
                                 
                                 ⁢ 
                                 
                                   T 
                                   c 
                                 
                               
                             
                             ) 
                           
                         
                       
                       + 
                       
                         
                           α 
                           j 
                           2 
                         
                         ⁢ 
                         
                           
                             X 
                             2 
                           
                           ⁡ 
                           
                             ( 
                             
                               n 
                               - 
                               
                                 
                                   τ 
                                   j 
                                 
                                 ⁢ 
                                 
                                   T 
                                   c 
                                 
                               
                             
                             ) 
                           
                         
                       
                     
                     ) 
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 3 
               
             
           
         
       
     
   
   Where τ j  is an integer and it indicates the delay of the j th  multi-path from the transmitter to the mobile. For simplicity of analysis we have assumed that the different multi-path delays are integer multiples of the chip width T c . The receiver structure does not significantly change when in reality the multi-path delays are not integer multiples of the chip widths. Now letting
 
 r   j ( n )=(α j   1   X   1 ( n−τ   j   T   c )+α j   2   X   2 ( n−τ   j   T   c ))  Equation 4
 
   and substituting Equation 4 into equation 3 one gets; 
   
     
       
         
           
             
               
                 
                   r 
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       j 
                       = 
                       1 
                     
                     L 
                   
                   ⁢ 
                   
                     
                       
                         r 
                         j 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     . 
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 5 
               
             
           
         
       
     
   
   For multiple receive antenna systems, such as a two receive antenna systems, MTA-MPIC can also differentiate between the other users being transmitted on the two transmit antennas. Thus, for two receive antennas, MTA-MPIC can also perform a MIMO type interference cancellation for the other users. 
   Referring now to  FIG. 2  there is shown a simple block diagram illustrating an embodiment of MTA-MPIC with one receive antenna and a HSDPA data user encoded for space-time transmit diversity. Initially, the signal is received at input  25  and each multi-path delayed signal is demodulated at unit  21 . Demodulator  21  can include long code removal and Walsh-Hadamard transform (WHT) despreading. Subsequently, each demodulated signal is received by the RAKE receiver  22 . The RAKE receiver  22  computes channel estimates and performs space-time decoding. Following the space-time decoding, at  230 , channel normalization is performed with respect to the HSDPA signals, and data decisions are made. Next, reconstructed interference signals corresponding to the space-time encoded HSDPA signals are generated by interference regenerator  24  in response to the decisions made at  23 . These interference signals  26  are combined with the received signal at a summing node  25  for interference cancellation. This procedure of demodulating the signal, rake, decision and canceling the multi-path interference can be repeated multiple times. Typical applications will do 2–3 iterations of the above procedure. 
   Referring now to  FIG. 3  there is shown a more detailed block diagram illustrating MTA-MPIC for the case of one receive antenna. The HSDPA signal is received from the antenna at input  11  and demodulated at  21 . Each multi-path finger for the HSDPA signal is delayed by an appropriate respective offset for synchronization by delay units  320  in which the received signal is delayed on the j th  path. Subsequently, any long coding is removed by units  321  for each finger. Following long code removal, the WHT despreader units  322  remove (i.e. despreads) the signal spreading codes. It is not necessary to have the Walsh code despreader for the individual codes. In general one could separate the long code despreader and the Walsh Hadamard despreader into two separate blocks. However, in order to reduce the complexity of the receiver WHT is employed to despread the Walsh codes, particularly when the number of HSDPA codes exceeds four. The combination of the long code despreader  321  and the WHT  322  effectively achieves the operation of, 
   
     
       
         
           
             
               
                 
                   
                     R 
                     j 
                     m 
                   
                   ⁡ 
                   
                     ( 
                     i 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       n 
                       = 
                       
                         iK 
                         + 
                         1 
                       
                     
                     
                       n 
                       = 
                       
                         
                           ( 
                           
                             i 
                             + 
                             1 
                           
                           ) 
                         
                         ⁢ 
                         K 
                       
                     
                   
                   ⁢ 
                   
                     
                       
                         C 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     ⁢ 
                     
                       r 
                       ⁡ 
                       
                         ( 
                         
                           n 
                           - 
                           
                             
                               τ 
                               j 
                             
                             ⁢ 
                             
                               T 
                               c 
                             
                           
                         
                         ) 
                       
                     
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 6 
               
             
           
         
       
     
   
   where the code C m (n) is multiplied to the received signal and summed over the spreading gain of the HSDPA channel, and i denotes a symbol. Note that even though the other release&#39;99 voice and other channels may have a spreading gain greater than the gain of the HSDPA channel they are also spread with the same spreading gain of the HSDPA channel only. This is done in order to reduce the complexity of the receiver of not requiring it to despread all the other channels with their large spreading gains. Without loss of generality it is assumed in equation 6 that i is an even integer. 
   Following demodulation  21 , the signals are received for a RAKE process  22 . Conventional conjugate (channel estimation) units  330  compute the channel estimate conjugate of the signals received from the despreader units  322 . Each WHT despreader  322  provides training channel information at  375  for conventional use by the corresponding channel estimation unit  330 . Each space-time decoder  337 , for path j, does the following space-time decoding for HSDPA code m&lt;{tilde over (M)}.
 
 Ŝ   1   m ( j )= R   j   m ( i )[α j   1 ]*+( R   j   m ( i+ 1))*α j   2 
 
 Ŝ   2   m ( j )=−( R   j   m ( i ))*α j   2   +R   j   m ( i+ 1)[α j   1 ]*  Equation 7
 
   The soft decisions for each of the symbols are now summed at adder  334  to produce; 
   
     
       
         
           
             
               
                 S 
                 ^ 
               
               1 
               m 
             
             = 
             
               
                 ∑ 
                 
                   j 
                   = 
                   1 
                 
                 L 
               
               ⁢ 
               
                 
                   
                     S 
                     ^ 
                   
                   1 
                   m 
                 
                 ⁡ 
                 
                   ( 
                   j 
                   ) 
                 
               
             
           
           ; 
           
             
               
                 S 
                 ^ 
               
               2 
               m 
             
             = 
             
               
                 ∑ 
                 
                   j 
                   = 
                   1 
                 
                 L 
               
               ⁢ 
               
                 
                   
                     S 
                     ^ 
                   
                   2 
                   m 
                 
                 ⁡ 
                 
                   ( 
                   j 
                   ) 
                 
               
             
           
         
       
     
   
   These summation results are input to a conventional channel normalization unit  336 , together with conventional channel estimate information from channel estimators  330 . Following channel normalization, a data decision is made at decision unit  23 . The decision unit  23  can either be a soft decision or a hard decision unit. The soft decision unit employs the soft decisions in the equation given above to generate the interference. The hard decision unit makes a hard decision on the bits of whether they are +/−1, using the soft decision as the input, depending upon whether it is a QPSK modulation or a 16 QAM modulation or some other modulation. Next, a reconstructed interference signal is generated at  24  for each finger beginning with HSDPA space-time encoding by encoding unit  342 . Subsequently, the signal is respread using WHT by unit  343  and multiplied by the long code by unit  344 . The signal is then delayed by the respective finger delays at delay units  346 . Each respective finger signal is next multiplied at  348  by the associated HSDPA channel estimate information received from channel estimate units  330 . Similar to equations 1 and 2, the estimated symbols for the HSDPA users are now ST encoded and respread, and the result is: 
   
     
       
         
           
             
               
                 
                   
                     
                       
                         X 
                         ^ 
                       
                       1 
                     
                     ⁡ 
                     
                       ( 
                       n 
                       ) 
                     
                   
                   = 
                   
                     
                       ∑ 
                       
                         m 
                         = 
                         1 
                       
                       
                         M 
                         ~ 
                       
                     
                     ⁢ 
                     
                       
                         
                           C 
                           m 
                         
                         ⁡ 
                         
                           ( 
                           n 
                           ) 
                         
                       
                       ⁢ 
                       
                         
                           
                             S 
                             ^ 
                           
                           1 
                           m 
                         
                         ⁡ 
                         
                           ( 
                           n 
                           ) 
                         
                       
                     
                   
                 
                 ⁢ 
                 
                   
 
                 
                 ⁢ 
                 and 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 8 
               
             
           
           
             
               
                 
                   
                     
                       X 
                       ^ 
                     
                     2 
                   
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       m 
                       = 
                       1 
                     
                     
                       M 
                       ~ 
                     
                   
                   ⁢ 
                   
                     
                       
                         C 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     ⁢ 
                     
                       
                         
                           S 
                           ^ 
                         
                         2 
                         m 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 9 
               
             
           
         
       
     
   
   A total regenerated interference is summed by adder  349 . For a given received finger the regenerated interference is now given by; 
   
     
       
         
           
             
               
                 
                   
                     I 
                     l 
                   
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       
                         j 
                         = 
                         1 
                       
                       , 
                       
                         j 
                         ≠ 
                         l 
                       
                     
                     L 
                   
                   ⁢ 
                   
                     ( 
                     
                       
                         
                           α 
                           j 
                           1 
                         
                         ⁢ 
                         
                           
                             
                               X 
                               ^ 
                             
                             1 
                           
                           ⁡ 
                           
                             ( 
                             
                               n 
                               - 
                               
                                 
                                   τ 
                                   j 
                                 
                                 ⁢ 
                                 
                                   T 
                                   c 
                                 
                               
                             
                             ) 
                           
                         
                       
                       + 
                       
                         
                           α 
                           j 
                           2 
                         
                         ⁢ 
                         
                           
                             
                               X 
                               ^ 
                             
                             2 
                           
                           ⁡ 
                           
                             ( 
                             
                               n 
                               - 
                               
                                 
                                   τ 
                                   j 
                                 
                                 ⁢ 
                                 
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                                   c 
                                 
                               
                             
                             ) 
                           
                         
                       
                     
                     ) 
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 10 
               
             
           
         
       
     
   
   As can be seen from the above equation, the interference for a given finger involves adding the estimate of the signals for the rest of the fingers. The operation of equation 10 can be realized by first adding all the signals at  349 , then subtracting the result from r(n) at  350 , and, lastly, summing the resultant of adder  350  with the respective regenerated interference signals at adders  352  to add back interference for each of the fingers to provide the individual finger signals. Thus, the adders at  349 ,  350  and  352  produce:
 
 r   l ( n )= r ( n )− I   l ( n )  Equation 11
 
   The above signal is now despread and space-time decoded again at  321 ,  322  and  337 , and the result is in turn used to regenerate a new estimate of the interference. The above procedure is repeated 2–3 times in some example embodiments to improve upon the interference estimate and the performance of the receiver. 
   An alternative embodiment for the one receive antenna case is illustrated in  FIG. 4 . In this embodiment, it is assumed that codes other than for HSDPA are being used (such as for voice users) and they are unknown to the receiver. Since the signal contribution of other users from the transmit antennas are considered to be independent signals, they cannot be discriminated individually using a single receive antenna without knowledge of the code and/or transmit scheme. This is why the  FIG. 3  embodiment treats the other users as interference and does not cancel them. The interference cancellation in  FIG. 3  only cancels the multi-path interference due to HSDPA channels. In the exemplary embodiment of  FIG. 4 , an interference estimate for other (e.g. voice) users is made by directly employing the despreader outputs. That is, instead of applying RAKE/space-time decoding to all the user signals, the other users are tapped before the full combining is determined, as shown at  400  in  FIG. 4 . 
   The approach illustrated in  FIG. 4  is similar to that illustrated in  FIG. 3  in all respects except that channels other than the HSDPA are tapped before the full combining is determined in the RAKE process  22 . The other users are tapped at the WHT despreaders  322  for each delay finger. This other user despreading can be done using those codes (of gain  16  in this example) that are not used by the HSDPA users. Additional despreading (not explicitly shown) can subsequently be applied. For example, voice users can have additional despreading by a factor of four, since voice users are typically employed with a spreading gain of 64 (i.e., 4×16=64). 
   Subsequently, the other user signals are respread at WHT respreaders  403  and an associated total regenerated interference is summed by adder  405  and then subtracted, along with the HSDPA regenerated interference, from the initial received signal by adder  350 . Lastly, the resultant of adder  350  is summed with the respective regenerated interference signals at adders  352  to add back interference from the other users and HSDPA for the fingers to provide the desired signals. Thus, for the other users, an interference estimate is made by directly employing the despreader outputs R j   m (i) (see equation 6). The regenerated interference for the voice users or the other channels is now given by; 
   
     
       
         
           
             
               
                 
                   
                     
                       I 
                       ~ 
                     
                     l 
                   
                   ⁡ 
                   
                     ( 
                     n 
                     ) 
                   
                 
                 = 
                 
                   
                     ∑ 
                     
                       
                         j 
                         = 
                         1 
                       
                       , 
                       
                         j 
                         ≠ 
                         l 
                       
                     
                     L 
                   
                   ⁢ 
                   
                     
                       ∑ 
                       
                         m 
                         = 
                         
                           
                             M 
                             ~ 
                           
                           + 
                           1 
                         
                       
                       M 
                     
                     ⁢ 
                     
                       
                         
                           C 
                           m 
                         
                         ⁡ 
                         
                           ( 
                           n 
                           ) 
                         
                       
                       ⁢ 
                       
                         
                           
                             R 
                             j 
                             m 
                           
                           ⁡ 
                           
                             ( 
                             n 
                             ) 
                           
                         
                         . 
                       
                     
                   
                 
               
             
             
               
                 Equation 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 12 
               
             
           
         
       
     
   
   Equation 12 can thus be used to calculate the interference from other users by using the despreader outputs directly for the other users. The regenerated interference is now subtracted out from the received signal (see  349 ,  350  and  352  of  FIG. 4 ), so the individual finger signals are given by;
 
 {tilde over (r)}   l ( n )= r ( n )− I   l ( n )− Ĩ   l ( n ).  Equation 13
 
   Referring now to  FIG. 6  there is illustrated an alternative embodiment, similar to  FIG. 4 , in which the other user interference is generated directly from despreader outputs, but with two receive antennas A 1  and A 2 . The approach illustrated in  FIG. 6  includes some duplicate structure  601  for operation with the second receive antenna (A 2 ). The duplicate structure  601  for antenna A 2  duplicates the following structure associated with antenna A 1 :  11 ; the  320 &#39;s; the  321 &#39;s; the  322 &#39;s; the  330 &#39;s; the  337 &#39;s; the  348 &#39;s;  349 ;  350 ; the  352 &#39;s; the  403 &#39;s;  405 ; and all interconnections therebetween. In duplicate structure  601 , as with antenna A 1 , channels other than HSDPA are tapped before the full combining is determined in the RAKE process. These other users are tapped at the WHT despreaders  322  for each antenna and each delay finger. Subsequently, the other user signals are respread at WHT respreaders  403  and an associated total regenerated interference is summed by adder  405  and then subtracted, along with the HSDPA regenerated interference, from the initial received signal by adder  350 . Lastly, the resultant of adder  350  is summed with the respective regenerated interference signals at adders  352  to add back interference from the other users and HSDPA for each delay finger to provide the desired signal. The outputs of all ST decoders  337  of  FIG. 6  are summed at  334 A, and the result is input to a conventional channel normalizer  336 A along with the channel estimate information from all channel estimators  330 . The output of channel normalizer  336 A is fed into the above-described processing path  23 ,  342 ,  343 ,  344 , and  346 &#39;s, and the outputs of the delays at  346  feed into the corresponding channel estimate multipliers  348  for both A 1  and A 2 . 
   Referring now to  FIG. 5  there are shown pertinent portions of further exemplary embodiments in which two receive antennas are used. The signal contribution of other voice users transmitted from multiple transmit antennas systems are again considered to be independent signals, however, they can be discriminated individually here because the presence of multiple receive antennas enables the individual reception of other voice users transmitted on multiple transmit antenna systems. 
   The approach illustrated in  FIG. 5  is similar to that illustrated in  FIG. 3 , but with two receive antennas A 1  and A 2 , a MIMO decoder  51  and a MIMO encoder  52 . In the  FIG. 5  embodiments, the following structure from  FIG. 3  is provided for each antenna (designated as A 1  and A 2 ):  11 ; the  320 &#39;s; the  321 &#39;s; the  322 &#39;s; the  330 &#39;s; the  337 &#39;s; the  348 &#39;s;  349 ;  350 ; the  352 &#39;s; and all interconnections therebetween. The channel normalizer  336 A in  FIG. 5  receives generally the same inputs as in  FIG. 6 . 
   In the embodiments of  FIG. 5 , the other user channels are processed differently. 
   Let
 
λ 1   m ( i )={ R   1   m ( i ),  R   2   m ( i ), . . . ,  R   L   m ( i )};  {tilde over (M)} +1 ≦m≦M   Equation 14
 
   indicate the ensemble of the despread signals for other users from antenna  1 . 
   Similarly define;
 
λ 2   m ( i )={ R   1   m ( i ),  R   2   m ( i ), . . . ,  R   L   m ( i )};  {tilde over (M)} +1 ≦m≦M   Equation 15
 
   which indicates the ensemble of the despread signals for other users from antenna  2 . 
   Then MIMO equalization and interference cancellation can be used to estimate the composite of the other user signals at spreading gain  16  on antennas  1  and  2  for users {tilde over (M)}+1≦m≦M. An exemplary MIMO decoder is illustrated in  FIG. 7 . 
   Conventional MIMO devices, namely minimum mean squared error (MMSE)/zero forcing (ZF) linear equalizers or with decision feedback, can be employed for the MIMO encoder and decoder of  FIG. 5 . The other user signals are tapped from despreaders  322  in generally the same manner as described above relative to  FIGS. 4 and 6 . These tapped signals are then input to the MIMO decoder  51 . The interference for the other users is then regenerated similar to the interference regeneration for HSDPA users, and is subtracted out from the received signal. 
   The decision unit  23 A produces decisions for both the HSDPA users (based on input from  336 A) and the other users (based on input from MIMO decoder  51 ). The HSDPA user decisions are processed the same as in  FIG. 3 . 
   The other user decisions are applied to MIMO encoder  52 , which in turn feeds a WHT respreader  343 A that applies respreading to both the HSDPA and other signals. From the output of WHT respreader  343 A, operations are the same as in  FIG. 3 , except the delay outputs from the  346 &#39;s are fed to the corresponding channel estimate multipliers  348  associated with both A 1  and A 2 . 
     FIG. 8  illustrates exemplary embodiments that utilize a weighting factor f (0≦f≦1) to weight the regenerated interference signals before they are combined with the received signal r(n). In some embodiments, the value of f increases with each iteration of interference regeneration. 
   Hence the interference cancellation of  FIG. 5 , in addition to removing the multi-path interference of the HSDPA channels, also removes the multi-path interference due to voice and other users. 
   The foregoing discussion for HSDPA is similarly applicable to other standards including; IX (CDMA 2000s), proposals for IX-EVDV (1 XTREME, L3QS), space-time spreading (STS) encoding in IX and proposals to IX-EVDV, WCDMA (release &#39;99 and others). Similarly, it can be extended to apply to more than 2 transmit and more than 2 receive antennas and for other transmit techniques such as double-STTD, double-STS and other multiple-input multiple-output (MIMO) techniques. 
   Although a preferred embodiment of the method and system of the present invention has been illustrated in the accompanied drawings and described in the foregoing Detailed Description, it is understood that the invention is not limited to the embodiments disclosed, but is capable of numerous rearrangements, modifications, and substitutions without departing from the spirit of the invention as set forth and defined by the following claims.