Patent Publication Number: US-6992632-B1

Title: Low profile polarization-diverse herringbone phased array

Description:
FIELD OF THE INVENTION 
     The present invention relates, in general, to an antenna and, more specifically, to a phased array antenna including multiple radiating elements arranged in a herringbone pattern. The phased array antenna operates over multi-octave bandwidths, subtends a wide field-of-view, and is capable of producing any desired polarization in space. 
     BACKGROUND OF THE INVENTION 
     Modern phased array systems are required to operate over wide frequency bandwidths with a single radiating aperture. In such broad band environments, processing functions that have previously been performed by individual antennas, need now to be performed by a single phased array. 
     A critical parameter of many RF signals is their polarization, requiring an array to respond to any linear, circular or elliptical polarization. This in the art is known as polarization diversity or polarization agility. Antenna polarization agility may be achieved with orthogonally disposed pairs of radiating elements that are electronically processed via a vector controller. Such a vector controller is described by Mohuchy in U.S. Pat. No. 5,933,108, issued Aug. 3, 1999 and is entitled “Gallium Arsenide Based Vector Controller for Microwave Circuits”, the disclosure of which is incorporated herein by reference. Polarization agility of a phased array is in much demand. 
     Significant advances in broadband solid-state power generation has also placed a new emphasis on phased arrays to efficiently combine the power of individual devices into high-power transmissions by exploiting a magnification property known as the “array factor”. Commensurate with this trend, demands for high transmitted effective radiated power (ERP) have increased. In addition, operating frequency range has been lowered into the HF/VHF region. 
     Dimensions of an antenna are inversely proportional to its operating frequency and wavelength, typically measured in tens of feet at HF/VHF frequencies. Consequently, size and weight of low-frequency antenna systems are of concern. This concern is particularly acute in mobile installations on aircraft, ground vehicles and even ships. To circumvent these limitations, shortened and inefficient antennas have been developed, which produced undesirable radiation performance and caused significant secondary inefficiencies in power and heat generation. More efficient radiators have been developed, such as Log-Periodic or Yagi arrays, which require considerable volume and are useable with the so called “big-bottle” transmitters. These radiators have limitations in broadband array applications, due to their element size that is incompatible with grating lobe suppression. 
     Conventional efforts on size reduction primarily addresses the microwave frequency region. A representative effort is disclosed by Wang et al. in U.S. Pat. No. 5,589,842, issued Dec. 31, 1996, which is entitled “Compact Microstrip Antenna with Magnetic Substrate”. Wang et al. disclose different planar radiators which typically are cavity backed for deployment on metallic surfaces. Although Wang et al. deal with arraying elements in circular disposition, they do not deal with any limitations brought about by grating lobe issues. 
     Dempsey et al. in U.S. Pat. No. 5,563,616, issued Oct. 8, 1996, which is entitled “Antenna Design using a High Index Low Loss Material”, disclose a high index of refraction medium having high matched values of relative permeability and relative permittivity. The Dempsey et al. approach favors the VHF frequency region. Applying their approach at HF frequencies, however, when deploying a polarization-diverse phased array, results in an antenna depth that far exceeds the space availability on most mobile platforms. 
     A need exists to drastically reduce the size of HF/VHF radiating elements, both in surface area and in depth. A need also exists to further decrease the element surface area of an array, while improving impedance characteristics of the array. Yet another need exists to improve the polarization capability of the array at HF/VHF frequencies. The present invention addresses these needs. 
     SUMMARY OF THE INVENTION 
     To meet this and other needs, and in view of its purposes, the present invention provides a phased array antenna including a plurality of radiating elements arranged as orthogonal pairs in a herringbone pattern. Each radiating element includes multiple microstrips disposed conformally on a planar substrate. Each radiating element includes a dipole formed as a pair of dipole microstrips extending from a pair of launch points. Each dipole microstrip extends between one launch point of the pair of launch points and a top loading microstrip, which provides a capacitive load to the dipole. The top loading microstrip extends between parallel microstrips, which provide an additional capacitive load to the dipole. 
     The multiple microstrips of the present invention are disposed approximately one-quarter wavelength above a ground plane. The planar substrate of the present invention is mounted on a composite substrate having a permittivity and permeability matched at a mid-band frequency of operation to achieve an impedance of approximately 377 ohms. The composite substrate is approximately 1/16 of a wavelength in thickness. The composite substrate includes an effective dielectric constant of approximately 10. 
     The composite substrate of the invention is mounted on a dielectric substrate having a dielectric constant value of approximately 98. The dielectric substrate is approximately 3/16 of a wavelength in thickness. Both the dielectric substrate and the composite substrate have an approximate thickness of ¼ of a wavelength and yield an approximate thickness reduction ratio of 6.6 to 1. 
     Another embodiment of the invention provides an antenna system including a phased array formed of a plurality of radiating elements arranged in a herringbone pattern, wherein the radiating elements are formed of multiple microstrips disposed conformally on a planar substrate. A transmit/receive network is connected to the radiating elements for varying the amplitude and phase of a transmitted signal. The transmit/receive network includes a receiver for determining direction and phase of a received signal, and a processor for controlling the amplitude and phase of the transmitted signal based on the direction and phase of the received signal. The transmit/receive network includes an array of modular transmitters for exciting a corresponding array of the radiating elements. 
     Yet another embodiment of the invention provides a method of making a phased array antenna. The method includes the steps of: (a) conformally forming multiple microstrips on a planar substrate, (b) arranging the multiple microstrips in a herringbone pattern, and (c) placing the multiple microstrips of the planar substrate approximately one quarter of a wavelength above a ground plane. The method also includes the step of: placing a composite substrate and a dielectric substrate between the planar substrate and the ground plane, wherein the composite substrate has an effective dielectric constant of approximately 10 and the dielectric substrate has an effective dielectric constant of approximately 98. The composite substrate is made approximately 1/16 of a wavelength in thickness, and the dielectric substrate is made approximately 3/16 of a wavelength in thickness. 
     It is understood that the foregoing general description and the following detailed description are exemplary, but are not restrictive, of the invention. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
       The invention is best understood from the following detailed description when read in connection with the accompanying drawing. Included in the drawing are the following figures: 
         FIG. 1  is a perspective view of multiple radiating elements configured in a herringbone pattern that are conformally mounted as microstrips on a multilayer substrate to form a planar phased array, according to an embodiment of the present invention; 
         FIG. 2  is a cross-sectional view along a central portion of a pair of radiating elements, or a pair of radiating dipoles, of the planar phased array shown in  FIG. 1 , in accordance with an embodiment of the invention; 
         FIG. 3  is a block diagram showing a transmit/receive network for use with the planar phased array of  FIG. 1 , according to an embodiment of the present invention; 
         FIG. 4  is a block diagram showing another transmit/receive network for use with the planar phased array of  FIG. 1 , according to an embodiment of the present invention; and 
         FIG. 5  is a block diagram showing still another transmit/receive network for use with the planar phased array of  FIG. 1 , according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to  FIG. 1 , there is shown a partial perspective view of a phased array antenna, in accordance with an embodiment of the invention. As shown, phased array antenna  6  includes multiple radiating elements  8 , which are arranged as orthogonal pairs in a herringbone pattern. Each orthogonal pair of radiating elements includes multiple microstrips that are formed conformally on a thin substrate. The substrate is generally designated as  17 . The orthogonal pairs of radiating elements  8  are positioned 45 degrees relative to a scan axis of the phased array antenna. 
     Each radiating element  8  includes a dipole formed from a pair of dipole microstrips, generally designated as  10   a  and  10   b . Each dipole microstrip of the pair of dipole microstrips extends from launch point  13  or launch point  14 . In this manner, each dipole is excited or fed in a balanced mode, at feed points or launch points  13  and  14 . For discussion purpose only,  FIG. 1  shows  8  radiating elements or  8  dipoles, each having launch points  13  and  14 . The phased array antenna may include more or less than 8 dipoles. 
     Dipole microstrip  10   a , for example, extends between launch point  13  and top loading microstrip  11 , the latter providing additional capacitance and a lowered Q for broadening the operational bandwidth of the dipole. Similarly, dipole microstrip  10   b  extends between launch point  14  and another top loading microstrip (not labeled). Top loading microstrip  11  extends between parallel microstrips  12   a  and  12   b , which are parallel to dipole microstrip  10   a . Parallel microstrips  12   a  and  12   b  provide additional capacitance and effective line stretching at lower frequencies of operation. 
     Parallel microstrips  12   a  and  12   b  may be tightly coupled to adjacent radiating elements. For example, as shown in  FIG. 1 , parallel microstrip  12   b  is tightly coupled to parallel microstrip  12   c , which forms one of another set of parallel microstrips of an adjacent radiating element (not separately labeled). The parallel microstrips may be meandered or interleaved for further coupling enhancement. Due to the tight mutual coupling, each dipole is effectively electrically longer at the lower end of the frequency band. The planar topography of the microstrips, shown in  FIG. 1 , and their tight coupling, in essence, forms a current sheet across the antenna array and significantly increases the operational bandwidth of the array. 
     The microstrips forming multiple radiating elements  8  may be produced by etching a conventional thin microstrip substrate  17 . Such microstrip substrate may be, for example, a Rogers 4000 series substrate, which is amenable to conventional processing methods. 
     As best shown in  FIG. 2 , along a cross-sectional view of dipole microstrips  10   a ,  10   b ,  10   c  and  10   d  of  FIG. 1 , the dipole microstrips are formed on thin substrate  17 . It will be appreciated that the dipole microstrips, as well as the top loading microstrips and the parallel microstrips, may be formed by etching standard thin substrate  17  for low power applications. If the radiated power is excessive, however, all the microstrips may be formed from solid metal strips. 
     The planar microstrips of the invention may typically be placed approximately one-quarter of a wavelength (λ/4) above ground plane  20 . The ground plane may be the metallic surface of a vehicle. In this manner, reflected backscatter energy from the metallic surface of the vehicle, or ground plane  20 , may recombine in-phase with the radiated signal from the dipole microstrips. Significant departure from this condition not only causes signal degradation, but in the limit, may entirely cancel radiation in the desired direction and may generate an undesirable surface wave. It is important, therefore, to form the electrical quarter-wave condition with minimum physical depth over a broad bandwidth. 
     The inventive compact microstrips  10   a ,  10   b ,  10   c  and  10   d , as well as  11  and  12   a  and  12   b  of dipoles  8   a  and  8   b  are typically etched on thin substrate  17 . This substrate, in turn, is mounted on composite substrate  18 , which may be fabricated in accordance with U.S. Pat. No. 5,563,616. Composite substrate  18  includes values of relative permittivity and relative permeability that are matched at the mid-band frequency of operation and achieves a free-space characteristic impedance of approximately 377 ohms. This impedance minimizes transitional reflective losses. The composite substrate forms a layer that is approximately 1/16 of a wavelength thick and presents an effective dielectric constant of approximately 10. Higher dielectric-constant materials may be impractical, because of the difficulty in providing a low-loss magnetic equivalent. 
     The next layer, designated as dielectric substrate  19 , includes dielectric material with a dielectric constant value of 98. Dielectric substrate  19  is approximately 3/16 wavelengths thick. The combined substrates, namely composite substrate  18  and dielectric substrate  19 , form a thickness of approximately ¼ of a wavelength. This combination yields an approximate depth reduction ratio of 6.6. Thus, for example, at 1 GHz the combined substrate thickness may be reduced from 2.95 inches to 0.45 inches, and at 100 MHz the thickness may be 4.5 inches instead of 29.5 inches. 
     Launch points  13  and  14  of dipole microstrips  10   a  and  10   b  are connected, respectively, to transmission feed lines  21  and  22 , as shown in  FIG. 2 . Similarly, the launch points (not labeled) of dipole microstrips  10   c  and  10   d  are connected, respectively, to transmission feed lines  23  and  24 . Center conductors  21   a  and  22   a  of transmission feed lines  21  and  22  are connected, respectively, to microstrips  10   a  and  10   b  at launch points  13  and  14 . Similarly, center conductors  23   a  and  24   a  of transmission feed lines  23  and  24  are connected, respectively, to microstrips  10   c  and  10   d  at their respective launch points (not labeled). 
     It will be appreciated that the outer ground conductors (not labeled) of transmission feed lines  21 ,  22 ,  23  and  24  may be electrically connected to ground plane  20 . The center conductors  21   a ,  22   a ,  23   a  and  24   a  pass through via holes (not labeled) in layers  18 ,  19  and  20  for eventual connections to dipole microstrips  10   a ,  10   b ,  10   c  and  10   d , respectively. 
     It will be understood that transmission feed lines  21 ,  22 ,  23  and  24  provide an interface to dipole microstrips  10   a ,  10   b ,  10   c  and  10   d  in pairs to receive/transmit channels for RF processing, as will be described later. These transmission feed lines may be either individual coaxial cables or twin-wire structures, as described in co-pending patent application Ser. No. 10/323,261 by the same inventor. Co-pending patent application Ser. No. 10/323,261 is incorporated herein by reference in its entirety, particularly with respect to its descriptions of twin-wire transmission lines. 
     Referring again to  FIG. 1 , phased array antenna  6  is shown as having four orthogonal pairs of radiating elements  8 . Although four orthogonal pairs of radiating elements are shown, the invention may be configured to include more or less orthogonal pairs than those four orthogonal pairs shown in  FIG. 1 . Moreover, generally each orthogonal pair of radiating elements includes two dipoles. Each dipole is generally excited in a balanced mode, at feed points or launch points  13  and  14 . 
     Exemplary approximate dimensions of the planar microstrips shown in  FIG. 1  will now be provided. As shown, the width of each dipole microstrip is 0.2 λ. The width of each parallel microstrip is 0.04 λ. The horizontal distance between one launch point of a dipole and another launch point of a horizontally adjacent dipole is 0.5 λ. The vertical distance between two opposing top loading microstrips of a dipole is 0.5 λ. The horizontal length of a top loading microstrip, when spanning between two parallel microstrips, is 0.49 λ. 
     A discussion will now be directed to RF networks for driving the phased array antenna of  FIG. 1 , as exemplified in  FIGS. 3–5 . Referring first to  FIG. 3 , there is shown a transmit/receive (T/R) network for exciting each dipole of the phased array antenna. The T/R network of  FIG. 3  is generally designated as  30  and may be modularly arranged, as shown. Transmit RF input port  38  is divided, at node  38   a , into multiple array columns, designated  45   a  through  45   n . Each array column corresponds to an orthogonal pair of radiating elements (only  8   a  and  8   b  are shown). As an example, in one embodiment of the invention, the four orthogonal pairs of radiating elements of  FIG. 1  may require four corresponding array columns  45   a ,  45   b    45   c  and  45   d . Since each array column includes similar modular components, only the components between array column  45   a  and radiating elements  8   a  and  8   b  (or dipoles  8   a  and  8   b ) will be described below. 
     Each array column, or more specifically array column  45   a  includes two paths, one path for each dipole. A first path includes variable attenuator  37   a  and variable phase shifter  36   a  connected in series. The attenuator may be used for controlling the transmitted polarization and channel alignment, if necessary. The phase shifter may be used for dual purposes. First, the phase shifter may provide appropriate phase setting for polarization. Second, the phase shifter may provide array beam steering. As will be explained, processor  44  controls both the phase setting of the phase shifter and the attenuation setting of the attenuator. 
     As shown, a power divider at node  35   a  splits the transmitted signal into two power amplifiers  34   a  and  34   b . In this manner, each power amplifier energizes a respective feed point of dipole  8   a . This is a preferred method of powering the phased array antenna instead of using a single amplifier  51  (shown in  FIG. 4 ). The advantage is derived from the separation of the two amplifiers results in a lower heat density. 
     The output signal of power amplifier  34   a , after passing through circulator  33   a  and 180 degree phase shifter  32   a , excites dipole  8   b . The output signal of power amplifier  34   b  passes through circulator  33   b  and then directly excites dipole  8   b . The two signals are 180 degrees out-of-phase, which assures proper phase excitation at launch points  13  and  14  ( FIG. 1 ) of dipole  8   b.    
     The second path includes variable attenuator  37   b  and variable phase shifter  36   b  connected in series. The attenuator may be used for controlling the transmitted polarization and channel alignment, if necessary. The phase shifter may be used for providing appropriate phase setting for polarization and array beam steering. Processor  44  may control both the phase shifter and the attenuator. 
     A power divider at node  35   b  splits the transmitted signal into two power amplifiers  34   c  and  34   d . In this manner, each power amplifier energizes each feed point of dipole  8   a . The output signal of power amplifier  34   c  passes through circulator  33   c  and 180 degree phase shifter  32   b  and then excites dipole  8   a . The output signal of power amplifier  34   d  passes through circulator  33   d  and then directly excites dipole  8   a . The two signals are 180 degrees out-of-phase which assures proper phase excitation of launch points  13  and  14  ( FIG. 1 ) of dipole  8   a.    
     In the example shown in  FIG. 3 , transmit module  31  includes attenuators  37   a  and  37   b , phase shifters  36   a  and  36   b , power dividers  35   a  and  35   b , power amplifiers  34   a ,  34   b ,  34   c  and  34   d . Although not shown, a similar transmit module  31  may be provided for the other array columns, such as  45   b ,  45   c ,  45   d , etc. 
     Operation of T/R network  30  in the receiving mode, as shown in  FIG. 3 , will now be described. The received signals are sampled by each of the planar compact dipoles  8  of phased array antenna  6 , for example dipoles  8   a  and  8   b . The received signals proceed toward direction finder (DF) receiver  42  and polarization receiver  43  for analysis. The received signals from the compact dipoles are separately combined by combiner  40   a  and combiner  41   a . Combiner  40   a  combines the signals in receive array columns  46   a ,  46   b ,  46   c , etc. and outputs the combined signals onto line  40  for input into DF receiver  42  and polarization receiver  43  for analysis. Similarly, combiner  41   a  combines the signals in receive array columns  47   a ,  47   b ,  47   c , etc. and outputs the combined signals onto line  41  as input to DF receiver  42  and polarization receiver  43  for analysis. 
       FIG. 3  shows the receive paths between compact dipoles  8   a  and  8   b  and receive array columns  46   a  and  47   a , respectively. The other receive paths between the other compact dipoles and the other receive array columns are similar and have been omitted in  FIG. 3  for purpose of clarity. 
     Following the receive signals from the upper transmission line (launch points  13  and  14  of dipole  8   b  shown in  FIG. 1 ), one feed point requires a 180 degree phase shifter  32   a  to assure that the dipole excitation is properly phased. The two signal paths of dipole  8   b  pass through circulators  33   a  and  33   b , and are combined at receive array column  47   a . Similarly, the lower transmission line (launch points  13  and  14  of dipole  8   a  shown in  FIG. 1 ) has one feed point passing through a 180 degree phase shifter  32   b  and circulator  33   c , while the other feed point passes through circulator  33   d . The two signal paths of dipole  8   a  are combined at receive array column  46   a.    
     It will be appreciated that circulators  33   a ,  33   b ,  33   c  and  33   d  may not provide sufficient isolation between the received and transmitted signals and may require either a limiter or a switch, or both, immediately after each circulator. These signals are combined into lines  40  and  41  by combiners  40   a  and  41   a . Combiner  40   a  represents right slant-linear polarization and combiner  41   a  represents left-slant linear polarization. These two channels are processed in DF receiver  42  to determine direction of arrival of the signal under analysis and, simultaneously, are analyzed in polarization receiver  43  for polarization, using techniques described in U.S. Pat. No. 5,933,108. The derived results are handed over to system processor  44  for controlling the transmit channels. The polarization techniques described in U.S. Pat. No. 5,933,108, are incorporated herein by reference in their entirety. As described therein, a vector controller includes receive ports for determining the incoming polarization of a received signal, and includes output ports for controlling the amplitude and phase of a transmitted signal. 
     It will be appreciated that the phased array antenna, when processed via a polarization control network, such as network  30 , has full polarization capability. The polarization may be controlled by open-loop methods or closed loop, adaptive methods. The closed loop, adaptive methods may encompass all linear polarizations and right or left hand circular polarizations. Open loop methods, however, may typically include six polarizations, namely vertical, horizontal, left slant-linear, right slant-linear, right-hand circular and left-hand circular. 
     Referring next to  FIG. 4 , there is shown an alternative transmit/receive (T/R) network, generally designated as  55 . Network  55  is similar to T/R network  30  shown in  FIG. 3 . The receive paths are similar to the receive paths of T/R network and are not shown in  FIG. 4 . Only the transmit paths between array column  45   a  and dipoles  8   a  and  8   b  are shown in  FIG. 4 . 
     A difference between T/R network  55  and T/R network  30  will now be described by referring to transmit module  56  of  FIG. 4 . Shown as an example, transmit module  56  includes attenuators  37   a  and  37   b , phase shifters  36   a  and  36   b  and single power amplifiers  51   a  and  51   b . The output signal of amplifier  51   a  is split by power divider  50   a  and sent to circulators  33   a  and  33   b . Similarly, the output signal of amplifier  51   b  is split by power divider  50   b  and sent to circulators  33   c  and  33   d . The other components shown in  FIG. 4  are similar to corresponding components shown in  FIG. 3 . A disadvantage of using a single amplifier, such as  51   a  and  51   b , is that a single amplifier must dissipate more heat than two separate amplifiers, such as amplifiers  34   a ,  34   b  or amplifiers  34   c ,  34   d , used in transmit module  31  of  FIG. 3 . 
       FIG. 5  illustrates a T/R network having a separate transmit module and a separate receive module, generally designated as  70 . Receive module  76  may use the inventive phased array antenna or may use other sensors to provide DF and polarization inputs. As shown, receive module  76 , of which only one receive array column is shown, includes antenna dipoles  71  and  72  connected to DF/polarization receivers  73  and  74 . The outputs from DF/polarization receivers  73  and  74  are sent to processor  75 . 
     Processor  75  may control the amplitude and phase of transmit module  31 , as previously described. Transmit module  31  includes the same components as those shown in  FIG. 3 . Only one transmit module is shown. The other transmit modules have been omitted for clarity purpose. 
     It will be appreciated that a difference in T/R network  70 , as compared to T/R network  30 , is the elimination of the entire receive path including circulators  33   a ,  33   b ,  33   c  and  33   d.    
     It will be further appreciated that the receive array size (number of columns) is determined by sensitivity requirements and does not need to be equal to the size of the transmit array. 
     Although the invention is illustrated and described herein with reference to specific embodiments, the invention is not intended to be limited to the details shown. Rather, various modifications may be made in the details within the scope and range of equivalents of the claims and without departing from the invention.