Patent Publication Number: US-7224400-B2

Title: Sharp-tuned filter-combiner for combining adjacent TV channels

Description:
RELATED APPLICATIONS 
   This application claims the benefit of U.S. Provisional Application Ser. No. 60/195,238 filed Apr. 7, 2000 and U.S. Provisional Application Ser. No. 60/236,225 filed Sep. 28, 2000. 

   BACKGROUND AND FIELD OF THE INVENTION 
   The present invention relates to transmitting television signals and, more particularly, to combining adjacent television channels such as two adjacent digital signals (DTV) or the combining of an analog signal (NTSC) with an adjacent digital signal (DTV). 
   DESCRIPTION OF THE PRIOR ART 
   Television signals have traditionally been broadcast in an analog format known as NTSC. The Federal Communications Commission (FCC) is now permitting a new digital format known as DTV. The digital format is presently in operation and the FCC has provided a transitional period until the year 2006 during which the NTSC signals and the DTV signals will both be transmitted. Thus, a station will simulcast both an NTSC signal and a DTV signal. It is understood that the FCC has allocated frequency bands or channels wherein an NTSC signal will be adjacent a DTV signal and also that a DTV signal will be adjacent to another DTV signal. In the United States, the channels are all 6 MHz wide, whereas in other parts of the world the channels are 6-8 MHz wide. The discussion presented herein is specifically directed to channels that are 6 MHz wide although it is to be understood that the discussion may be similar for channels that are up to 8 MHz wide. 
   When a DTV allocation is one channel below an NTSC channel, this is referred to as the N−1 case. When the DTV allocation is one channel above an NTSC channel, this is referred to as the N+1 case. 
   To prevent adjacent channels from interfering with each other, it has been known in the past to employ bandpass filters and hybrid couplers, simply referred to herein as hybrids, to form bandpass filter-combiners. Bandpass filter-combiners are described in a four page article entitled, Adjacent Channel Combining for N+1 DTV Channel Allocations, in the ADC Technical News Publication, dated Jul. 30, 1998 published by ADC Telecommunications, Inc.  FIG. 1  herein is based on  FIG. 1  in that publication and presents a bandpass filter-combiner employing a pair of bandpass filters  10  and  12  interposed between hybrids  14  and  16 . One of the input ports of hybrid  14  is coupled to a reject load  18  and the other input receives an RF signal A. Another RF signal B is supplied to a port on filter  16 . The bandpass filters  10  and  12  are tuned to pass signal A but reflect signal B. Consequently, signal A enters the left side port of hybrid  14  and the signal splits into portions which pass through the bandpass filters  10  and  12  and thence into the hybrid  16  and combine with signal B. Signal B entered the hybrid and then split with portions reflecting off the bandpass filters  10  and  12  and returning into the hybrid to combine with signal A to provide a combined signal A+B at an output port of hybrid  16 . The article points out that the filter-combiner of  FIG. 1  is not practical for use in an N+1 case wherein the DTV channel is above the NTSC channel. This is because the aural carrier in the N+1 case is only 0.25 MHz away from the upper channel edge, and may have sidebands up to 120 kHz away. Thus, the guard band is very small. 
   The ADC article proposes a design for the N+1 case and this takes the form shown herein at  FIG. 2  which is based on  FIG. 5  in the ADC article.  FIG. 2  herein shows a constant impedance filter-combiner  20  in combination with a notch diplexer  22 . The constant impedance filter-combiner  20  is based on that illustrated herein at  FIG. 1  and, consequently, like components are identified with like character references. Signal A is replaced with a DTV signal and signal B is replaced with a VISUAL signal from an NTSC signal. The AURAL signal has been removed and, as will be seen, is re-inserted downstream. The bandpass filters  10  and  12  are tuned to pass the DTV signal and reflect the VISUAL signal. The output port will provide a combined signal including the DTV signal and the VISUAL signal. This signal is then supplied to the notched diplexer where the AURAL signal is re-inserted. The notched diplexer is tuned to reject or reflect the AURAL signal and to pass the combined DTV and VISUAL signals obtained from the filter-combiner  20 . The notched diplexer includes a left hybrid  24  having an input port for receiving the combined signal from the filter-combiner  20  and another input port connected to a load  26 . The notched diplexer includes a pair of aural notch filters  28  and  30  that extend from the left hybrid  24  to a right hybrid  32 . The AURAL signal is supplied to a port  34  of hybrid  32  and the AURAL signal is then split and portions reflect back from filters  28  and  30  so that the AURAL signal combines with the DTV and VISUAL signals and the combined signal appears at an output port  36  for application to an antenna  38 . It is to be noted that most notched diplexers also include −3.58 MHz traps in series with the AURAL notched filters, such as filters  28  and  30  in  FIG. 2 , to provide attenuation of unwanted color difference products in the lower sideband. For simplicity, such traps have not been included in  FIG. 2 . 
   By splitting the NTSC signal into its VISUAL and AURAL components, the prior art presented in  FIG. 2  requires that the AURAL signal be added down stream. Consequently, in order to handle the N+1 case, this type of prior art requires a circuit having three inputs for receiving the DTV, VISUAL, and AURAL signals. Additionally, this circuit requires the use of four hybrids  14 ,  16 ,  24  and  32 . It is desirable to reduce the cost of the equipment needed to handle an N+1 case. 
   Attention is now directed to  FIG. 3  which illustrates another prior art circuit for handling the N+1 case. This circuit includes only two hybrids  40  and  42  as opposed to the four hybrids employed in the circuit of  FIG. 2 . The left hybrid  40  has an input port connected to a load  41  and a second input port that receives a DTV signal by way of a power amplifier  44 . The DTV signal splits as it enters the hybrid  40  and a portion of it is passed by a DTV bandpass filter  46  and another portion is passed by a DTV bandpass filter  48 . These DTV portions are passed by AURAL notches  50  and  52  and then enter the right hybrid  42 . An NTSC signal that includes both AURAL and VISUAL components is supplied to a port on the right side of hybrid  42  and these signals split in the hybrid and the AURAL portions are reflected from the AURAL notches  50  and  52  and are passed back through the hybrid and recombined and are applied to an antenna  60 . The VISUAL portions of the NTSC signal pass through the AURAL notch cavities of the AURAL notches  50  and  52  and are reflected by the DTV bandpass filters  46  and  48  and return through the AURAL notches  50  and  52  and thence into the right hybrid  42  and combine with the DTV and AURAL signals and supplied to the antenna  60 . Whereas the circuit of  FIG. 3  only has two hybrids, it is to be noted that the NTSC VISUAL signal passes over the AURAL notches twice during the operation adding a heat burden to the AURAL cavities. Any temperature drifting will cause performance parameters to fall outside the equalization set up because there is no tracking adaptive system in NTSC. Also, it is to be noted that a notch diplexer type combiner as shown in  FIG. 3  results in reduced DTV to NTSC chroma isolation. Additionally, the AURAL notch causes equalization problems for the transmitter system because the notched bandwidth can cut back into the NTSC chroma region which causes additional high frequency video equalization adjustments. 
   From the foregoing discussion of the prior art circuits of  FIGS. 1 ,  2  and  3 , it is seen that an improvement is needed in the bandpass filters that are employed in these combiners. A more sharply tuned filter is needed that will pass the DTV signal for a particular channel while reflecting all other frequencies including an adjacent channel NTSC signal or an adjacent channel DTV signal. The filter should exhibit an amplitude response within a mandated mask (such as the FCC mandated mask). Such a sharply tuned filter can be used as a stand alone filter having deep levels of IMD sideband rejection at the channel edges for additional FCC mask suppression. This enhances transmitter performance so that the transmitter power may be increased substantially while meeting the requirements of the FCC mandated mask. 
   In addition to operating as a stand alone mask filter resulting in power enhancement, such sharply tuned filters may be employed in a constant impedance filter-combiner without the need for notched filters for the AURAL carrier signal and, hence, such a combiner may be employed in the N+1 case as well as in the N−1 case as well as for adjacent channel DTV signals. 
   SUMMARY OF THE INVENTION 
   In accordance with one aspect of the present invention, an apparatus is provided for combining television signals. The combiner has two inputs for respectively receiving an NTSC signal, including both VISUAL and AURAL components, and a DTV signal wherein the NTSC signal and the DTV signal are in adjacent channels with the DTV signal being of greater frequency than the NTSC signal. The combiner has sharply tuned filtering. The filtering is tuned for passing the DTV signal while reflecting the NTSC signal and such that the passed DTV signal and the reflected NTSC signal combine as a combined output signal. 
   In accordance with another aspect of the present invention, the combiner includes first and second hybrids and first and second sharp tuned filters. The filters are tuned to pass only the DTV signal while reflecting the NTSC signal. 
   In accordance with another aspect of the present invention, the signal combiner has two inputs for respectively receiving a first DTV signal and a second DTV signal wherein the TV signals are of different frequencies and in adjacent channels. The combiner has sharply tuned filtering such that the first DTV signal is passed while the second DTV signal is reflected. The passed and reflected signals combine to provide a combined output signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The foregoing and other objects and advantages of the present invention will become more readily apparent from the following as taken in conjunction with the accompanying drawings wherein: 
       FIG. 1  is a block diagram illustration of a bandpass filter-combiner in accordance with the prior art; 
       FIG. 2  is a block diagram illustration of another bandpass filter-combiner in combination with a notch diplexer in accordance with the prior art; 
       FIG. 3  is a notch type combiner also in accordance with the prior art; 
       FIG. 4  is a block diagram illustration of a sharp tuned filter constructed in accordance with the present invention and illustrated in combination with an amplifier and an antenna; 
       FIG. 5  is a graphical illustration of amplitude with respect to frequency and illustrating typical filter specifications for amplitude response and also including the FCC mask and the sharp tuned filter mask herein; 
       FIG. 6  is a graphical illustration of time in nanoseconds versus frequency and illustrating typical filter specifications with respect to group delay; 
       FIG. 7  illustrates an embodiment of the invention herein employing two sharply tuned filters and two hybrids employed for supplying an amplified DTV signal to an antenna for broadcasting; 
       FIG. 8  is a block diagram illustration of an embodiment of the invention herein for performing adjacent channel combining of an NTSC signal and a DTV signal in an N+1 case or an N−1 case; 
       FIG. 9  is a graphical illustration of amplitude with respect to frequency showing the operation of the embodiment in  FIG. 8  for adjacent NTSC and DTV channels in the N+1 case; 
       FIG. 10  is a graphical illustration of amplitude with respect to frequency showing the operation of the embodiment in  FIG. 8  for adjacent NTSC and DTV channels in the N−1 case; 
       FIG. 11  is similar to that of  FIG. 8  but employed for combining adjacent channel DTV signals; 
       FIG. 12  is a graphical illustration of amplitude with respect to frequency showing the operation of the circuitry in  FIG. 11  with respect to adjacent DTV channels; 
       FIG. 13  is another embodiment of the invention employing adaptive equalization RF sample feedback; 
       FIG. 14  is another embodiment of the invention similar to that as illustrated in  FIG. 13 , but including additional features; 
       FIG. 15  is a graphical illustration of amplitude with respect to frequency that show the lower adjacent channel response in an N+1 case; 
       FIG. 16  is a graphical illustration of amplitude with respect to frequency illustrating the DTV bandpass amplitude response; 
       FIG. 17  is a graphical illustration of time in nanoseconds with respect to frequency illustrating the group delay response of the sharp tuned filter; 
       FIG. 18  is a graphical illustration of amplitude with respect to frequency showing the video response of the filter herein; 
       FIG. 19  is a graphical illustration of time in nanoseconds with respect to frequency illustrating the video group delay of the sharply tuned filter the reduction in the equalization requirement of the transmitter; 
       FIG. 20  is a graphical illustration of amplitude with respect to frequency showing improved isolation; 
       FIG. 21  is a graphical illustration of amplitude with respect to frequency illustrating the monaural amplitude response of a sharp tuned filter; 
       FIG. 22  is a graphical illustration of amplitude with respect to frequency showing a stereo signal amplitude response of the filter herein; 
       FIG. 23  is a graphical illustration of time with respect to frequency showing a stereo signal group delay response of the filter herein; 
       FIG. 24  is a graphical illustration of amplitude with respect to frequency showing a typical transmitter spectral spread of a DTV signal; 
       FIG. 25  is a graphical illustration similar to that of  FIG. 24  but showing the 0.5 MHz FCC integrated power zone; 
       FIG. 26  is a graphical illustration similar to that of  FIGS. 23 and 24  but illustrating the operation of a transmitter employing a sharp tuned filter constructed in accordance with the invention herein; 
       FIG. 27  is similar to  FIGS. 24-26  but illustrating an enhanced high power operation of a transmitter without a sharp tuned filter and with a sharp tuned filter constructed in accordance with the filter described herein; 
       FIG. 28  is a block diagram similar to  FIG. 4  but including a feedback path and an exciter; 
       FIG. 29  is an elevational view illustrating a waveguide that may be employed in practicing the invention herein; 
       FIG. 30  is a view taken along line  30 - 30  looking in the direction of the arrows in  FIG. 29  and illustrating the configuration of the input port of the waveguide; 
       FIG. 31  is a view taken along line  31 - 31  looking in the direction of the arrows in  FIG. 29  illustrating an iris plate; 
       FIG. 32  is a view taken along line  32 - 32  looking in the direction of the arrows in  FIG. 29  illustrating another iris plate; 
       FIG. 33  is a view taken along line  33 - 33  looking in the direction of the arrows in  FIG. 29  and showing a still further iris plate; 
       FIG. 34  is an enlarged perspective view of the filter that also illustrates adjustment screws (probes); and 
       FIG. 35  is a sectional view showing iris plate  111 . 
   

   DESCRIPTION OF PREFERRED EMBODIMENTS 
   Reference is now made to  FIG. 4  which illustrates a sharp tuned filter (STF)  100  constructed in accordance with the present invention and which is illustrated as having a single input port  101  and a single output port  102 . The input port is illustrated as being connected to the output of an amplifier  104  which, for example, may receive and amplify a digital DTV signal for a particular channel, such as channel  10 . The DTV signal is applied to the sharp tuned filter  100  which is tuned to pass only signals in this channel, i.e. channel  10  in the example being described. The passed DTV signal is then provided at the output port  102  and forwarded to a load, such as an antenna  106  for broadcasting purposes. 
   It is to be understood that the load may take a form other than an antenna. The filter  100  may take various forms such as a length of coaxial cable or a wave guide. The structural configuration may take these or other forms so long as the filter complies with the specifications noted below and given with reference to the graphical illustrations in  FIGS. 5 and 6 . For example, the filter  100  may take the form of waveguide  101  shown in  FIGS. 29-33 . This waveguide is a hollow cylinder constructed of suitable conductive metal, such as aluminum, copper or steel. The waveguide may have multiple cavities, such as cavities  103 ,  105  and  107  defined by iris plates  111 ,  113 , and  115 . The iris plates and the three sections of the waveguide may be held together as with welding or bolt and nut arrangements. The waveguide has an input port  117  and an output port. The input port is of rectangular shape, as indicated at  FIG. 30 . The iris plates have iris openings of various shapes. For example, the iris opening in plate  111  is a cross  121  as shown in  FIG. 31 . The iris opening in plate  113  is a horizontal slot M 45  as shown in  FIG. 32 . The iris opening in plate  115  is a vertical slot M 60 . As best shown in  FIGS. 34 ,  35 , the waveguide  101  also has mode resonance screws i-vi and mode coupling screws M 12 , M 34  and M 56  that protrude into the cavities. These screws (or probes) are made of metal, such as copper, and extend into the cavities by various amounts, such as in the range from 0.25 inch to 0.50 inch for a circular waveguide having an inner diameter on the order of 20.0 inches. Note that cavity  103  is for mode resonance numbers  1 ,  2  and that cavity  105  is for mode resonance numbers  3 ,  4  and that cavity  107  is for mode resonance numbers  5 ,  6 . The cross  121  includes slot M 14  that couples resonances  1  and  4  and slot M 23  that couples resonances  2  and  3 . The horizontal slot M 45  couples resonances  4  and  5 . 
   The filter sections are realized thorough distributed capacitances and inductances through the use of the circular waveguide cavity sections separated by conductive iris plates. The size of the cavities is such that two modes (such as modes  1 ,  2  or  3 ,  4  or  5 ,  6 ) are allowed to propagate through the waveguide (as opposed to the propagation of only the dominant mode). 
   The filter is tuned during construction to comply with the specifications set forth in  FIGS. 5 and 6 . The tuning includes varying the length and inner diameter of the waveguide and varying the shape and size of the iris openings and the orientation of the openings relative to each other. The waveguide may be on the order of 6 feet in length and about 20 inches in inner diameter. Additional tuning during filter construction involves adjusting the screws (probes) i-vi and M 12 , M 34  and M 56 . For example a first adjustment may be made as to the depth that these screws protrude into the waveguide cavities. Then a test may be made by applying RF energy of the frequency channel of interest and then a determination is made with suitable meters (such as a spectrum analyzer) as to the filter response relative to the FCC mask. This step may be repeated until the response meets the specifications herein. The adjustment of the mode resonance screws i-vi change the resonant frequency in the associated cavity. The filter bandwidth is changed by adjusting the three mode coupling screws M 12 , M 34  and M 56 . Deeper screw penetration provides greater mode coupling. This increases filter bandwidth. The opposite adjustment decreases filter bandwidth. 
   In  FIG. 5  there is illustrated, in dotted lines, the FCC mask  110  for DTV signals sometimes known as the 8VSB standard or the 8VSB modulated RF signal. The Federal Communications Commission (FCC) has mandated that each television channel have a bandwidth of 6 MHz whether the channel be a DTV channel or an NTSC channel. The FCC mask  110  requires that all signals broadcasted have their power attenuated starting at frequencies no greater than ±3.5 MHz relative to the center frequency Fc of the assigned channel. The attenuation is complete. The FCC mask, as shown in  FIG. 5 , requires that the attenuation be continuous within the mask. The mask has left and right skirts  112  and  114  that extend in a linear fashion from mask edges  116  and  118  from the in-band power level  120  to −64 dB at ±9 MHz relative to the center frequency Fc. The in-band power level  120  will sometimes be referred to herein as the reference level. 
   The filter  100 , in accordance with the present invention, complies with and falls within the mandated FCC mask as is indicated herein by the solid line  130  representing the filter mask of filter  100 . This shows the amplitude response of the filter. The vertical dashed lines  140  and  142  represent the 6 MHz bandpass from −3.0 to +3.0 MHz relative to the center line frequency Fc that must be passed by the filter. Attenuation of signals beyond ±3.0 MHz up to about ±3.45 MHz, as indicated by the dashed lines  144  and  146 , is achieved by the filter  100 . This attenuation is uniform about the center frequency extending downward to an attenuated level as indicated by the horizontal dashed line  148  and this attenuated level is at about −1 to −18 dB from the in-band power level  120 . From this attenuated level, the amplitude response is further attenuated in a skirt like fashion to a level  150  of about −40 dB to −64 dB at ±9 MHz relative to the center line frequency Fc. 
   There may be some amplitude ripple at the in-band power level  120 , however, this should stay within a response window  152  and not exceed about −0.5 dB below the in-band power level  120 . Additionally, the insertion loss  154  should not exceed about −0.20 dB from the in-band power level  120 . 
   Reference is now made to  FIG. 6  which presents a graphical illustration of time with respect to frequency showing the group delay as represented by curve  200  within the 6 MHz bandpass as represented by vertical dashed lines  202  and  204  at −3.0 and +3.0 MHz relative to the center frequency Fc. Points A and B are taken at −2.69 and +2.69 MHz relative to the center frequency Fc as indicated by the vertical dashed lines  206  and  208 . These lines  206  and  208  intersect curve  200  at points A and B which are to be kept within 50 nanoseconds of each other. 
   The specifications of the filter  100  as presented in  FIGS. 5 and 6  and as discussed above have been presented relative to the standards in the United States wherein the FCC has allocated television channels as being 6.0 MHz wide. The European and other non-U.S. standards differ somewhat and, for example, the bandpass filter must be modified to pass frequencies on the order of 6 to 8.0 MHz which is the channel width or bandwidth in other parts of the world. Consequently, if the bandwidth of each channel is designated as being on the order of W MHz, then W may be 6 for the United States and 6 to 8 MHz for other parts of the world. 
   The filter described herein with respect to  FIGS. 4 ,  5  and  6  may be employed as a stand alone mask filter with deep levels of IMD sideband rejection on the channel edges. Such a stand alone mask filter is illustrated in  FIG. 4  and the operation is described herein in greater detail with reference to  FIG. 27 . Such a stand alone mask filter enhances transmitter performance by reducing the linearity requirements to meet the FCC mask and allows higher power transmitter operation for improved power output. As such the filter provides greater out-of-channel protection to guard against potential NTSC interference which, in many cases, could be the same NTSC station due to current adjacent channel allocations. 
   Additionally, the filter  100  as described relative to  FIGS. 4 ,  5  and  6 , may be employed in conjunction with another filter in a stand alone filter mask in the manner as set forth in  FIG. 7 . There, a pair of filters  100 A and  100 B, of identical construction to each other and to filter  100 , are interposed between a left hybrid  300  and a right hybrid  302 . The filters  100 A and  100 B are tuned to pass the frequency channel of the DTV signal. The left hybrid  300  has an input port  304  connected to a load  306  and a second input port  308  that receives an amplified DTV signal from an amplifier  310 . The DTV signal enters the hybrid  300  and is split into two paths, one extending through the filter  100 A and the other passing through filter  100 B. The two signals enter the right hybrid  302  where they combine and provide a DTV signal at the output port  312  for application to an antenna or load  314 . Another port on the right hybrid  302  is connected to a load  316 . As will be discussed in greater detail hereinafter with reference to  FIG. 27 , such a circuit will easily meet the FCC sideband suppression levels while allowing operation of the transmitter at a substantially higher power level. This will be discussed in greater detail hereinafter with reference to  FIG. 27 . 
   Attention is now directed to  FIG. 8  which illustrates a constant impedance filter-combiner  400  constructed in accordance with the present invention and employing a pair of filters  100 C and  100 D each constructed in the same manner as filter  100 . Each filter  100 C and  100 D is tuned to pass television RF signals within a particular channel while rejecting all other RF frequencies. In the example being presented, the mode of operation is for the N+1 case. Consequently, the DTV signal is from a channel of higher frequencies than the NTSC signal. For example, the DTV channel may be that for channel  10  (192 MHz to 198 MHz) and NTSC channel may be channel  9  (186 MHz to 192 MHz). In this example, both filters  100 C and  100 D are tuned to pass the DTV signal (channel  10 ) while rejecting all other RF frequencies. The DTV signal is supplied to a power amplifier  402  and, thence, to an input port  404  at the left side of a hybrid  406 . Another input port  408  of hybrid  406  is connected to a reject load  410 . The DTV signal enters the hybrid  406  at the input port  404  and then is split into two portions which are respectively passed by the filters  100 C and  100 D and enter a right hybrid  414  and are recombined and are supplied to an output port  416 . The NTSC signal, including both audio and video components, is supplied to a port  418  on the right side of hybrid  414 . The signal is then split in the hybrid  414  and portions exit from the left side of hybrid  414  and are reflected from the filters  100 C and  100 D and reenter the hybrid  414  and recombine along with the recombined DTV signal and are provided at output port  416  and applied to a load, such as antenna  420 . The hybrids in  FIGS. 7 and 8  may be zero degree or ninety degree hybrids. 
   The filter-combiner presented in  FIG. 8  requires only two hybrids and only two input ports ( 404  and  418 ) as opposed to the construction of the prior art filter-combiner in  FIG. 2  that requires four hybrids and three input ports. Moreover, by providing such sharp tuned filters as filter  100 C and  100 D, there is no need to provide AURAL notches such as AURAL notches  50  and  52  in the prior art filter-combiner of  FIG. 3 . 
     FIGS. 9 and 10  illustrate radiation patterns of amplitude with respect to frequency for the N+1 case and the N−1 case, respectively. Thus, in the N+1 case as shown in  FIG. 9 , the NTSC signal is in the lower frequency channel, such as channel  9 , and the DTV signal is in the adjacent higher frequency channel. This is a spectrum analyzer presentation and it shows low DTV power which is normal. 
   The filter-combiner of  FIG. 8  may also be employed in the N−1 case, as shown in  FIG. 10 , such as wherein the DTV signal is the lower channel, such as channel  10 , and the NTSC signal is in a higher channel, such as channel  11  (198 MHz to 204 MHz). Both  FIGS. 9 and 10  show the radiation patterns as measured and each division in the vertical direction represents 10 dB and each division in the horizontal direction represents 2 MHz. 
   Reference is now made to  FIG. 11  which illustrates a combiner  400 ′ which is virtually identical to combiner  400  and consequently like components are identified with like character references. The significant difference is that this combiner combines two DTV signals, one represented as D 1  which is applied through amplifier  402  to the input port  404  of the left hybrid  406  in the same manner as discussed hereinbefore with reference to  FIG. 8 . In the case of  FIG. 11  the DTV signal D 1  passes through the hybrids  406  and  414  and the filters  100 C and  100 D as indicated by the dashed lines in the same manner as the discussion presented above with reference to  FIG. 8 . The filters  100 C and  100 D are tuned to pass only the DTV signals in the channel for the digital signal D 1  (channel  10  192 MHz to 198 MHz) and reject all other frequencies. The second digital signal D 2  may be from an adjacent channel which is either higher or lower than that of channel  10 . In the event that it is of a higher frequency then it will be from channel  11 . Since the filters  100 C and  100 D are tuned to reject such frequencies, the signal D 2  will be split as it enters the right hybrid  414  and the split portions are reflected from filters  100 C and  100 D and pass back through the hybrid, as indicated by the dotted lines and combine with the digital signal D 1  for application to the antenna or load  420 . An amplitude versus frequency spectral plot for typical adjacent DTV channels is illustrated in  FIG. 12 . 
   Reference is now made to  FIG. 13  which is similar to  FIGS. 8  or  11  in that it employs a filter-combiner  400 ′ and may operate in an N+1, N−1 or a D+D configuration. As illustrated, a DTV transmitter  423  with an exciter  419  supplies a digital signal D 1  to input port  404  and either an NTSC signal or a second DTV signal is supplied to input port  418  and the output is obtained from output port  416  and supplied to an antenna  420 . An adaptive feedback path  417  provides feedback to the DTV exciter  419 . A digital filter  421  at the input to the exciter  419  removes any interference that may be obtained from the NTSC signal or the second DTV signal. The exciter includes pre-correction circuitry that pre-corrects the information signal supplied to antenna to correct for any distortion caused by the filters in the combiner-filter  400 ′. The pre-correction may be for both non-linear and linear distortions. Preferably the correction is for at least any linear distortion introduced by the filters in the filter-combiner  400 ′. As shown, the correction may be adaptive with a feedback path. The correction may also be non-adaptive, without the feedback path. 
   Reference is now made to  FIG. 14  which illustrates another embodiment of the invention similar to that with reference to  FIG. 13  and which may be used when transmitting two DTV signals referred to herein as signals D 1  and D 2 . For purposes of illustration assume that D 1  is the lower frequency signal and D 2  is the higher frequency signal. This embodiment employs a pair of exciters  500  and  502  which supply the D 1  and D 2  signals to a pair of amplifiers  564  and  506 . The amplified D 1  and D 2  signals are supplied to a filter-combiner, such as combiner  400 ′ from  FIG. 11 . The D 1  signal is supplied to input port  404  and the D 2  signal is applied through the filter  508  and thence to input port  418 . The filter  508  is employed for passing the D 2  signal and rejecting all other RF signals and then the signal is applied to the filter-combiner. The output signal which includes the combined D 1  and D 2  signals is supplied from the output port  416  to the antenna  420 . In this embodiment, a sample is taken of the output signals D 1  and D 2  and fed back to the exciters  500  and  502  with the use of a signal splitter  510 . Each of the exciters is provided with a filter to remove interference from the non-related exciter. Thus exciter  500  is provided with a filter  512  to remove any interference from signal D 2 . Exciter  502  is provided with a filter  514  to remove any interference from signal D 1 . Both exciters  500  and  502  include pre-correction circuitry that pre-corrects the information signal supplied to antenna to correct for any distortion caused by the filters in the combiner-filter  400 ′. The pre-correction may be for both non-linear and linear distortions. Preferably the correction is for at least any linear distortion introduced by the filters in the filter-combiner  400 ′. As shown, the correction may be adaptive with a feedback path. The correction may also be non-adaptive, without the feedback path. 
     FIGS. 15 through 23  illustrate many of the features of the response characteristics of the filter described herein. These are specifically related to operation of the embodiment illustrated in  FIG. 8  when operating with adjacent channels in an N+1 case. 
     FIG. 15  illustrates the response curve between the NTSC and the DTV passband zones. The sharp tuned characteristic is evident in the very close frequency spacing. Note the video frequency points marked as A, B and C. Point A is the chroma carrier at 3.58 MHz above the visual carrier, point B is the video upper band edge at 4.2 MHz above the visual carrier which is attenuated only −0.4 dB and point C is the reference visual carrier and shows little attenuation at the upper sideband frequencies around the chroma zone. 
     FIG. 16  illustrates the DTV bandpass showing useful in-band zone covering an inner span of 5.9 MHz. The amplitude response just beyond 5.9 MHz shows a rapid down turn in the response as the effect of the filter becomes apparent. This is a very sharp roll off and is beneficial for removing out of band IMD distortion products. 
     FIG. 17  is a graphical illustration of time with respect to frequency illustrating a typical group delay response of the DTV bandpass of the filter herein. 
     FIG. 18  is a graphical illustration of amplitude with respect to frequency showing the video response of the filter relative to the lower adjacent channel (NTSC) and shows little attenuation at the upper sideband frequencies around the chroma zone. 
     FIG. 19  is a graphical illustration of time with respect to frequency showing the group delay characteristics of the lower adjacent channel. 
     FIG. 20  is a graphical illustration of amplitude with respect to frequency and shows the improved isolation needed for satisfactorily combining NTSC and DTV signals. 
     FIGS. 21 ,  22  and  23  illustrate audio performance aspects of the sharply tuned filter. 
     FIG. 21  is a graphical illustration of amplitude versus frequency and the monaural bandwidth and which is expanded from that illustrated herein at  FIG. 15 . 
     FIG. 22  shows the stereo bandwidth. 
     FIG. 23  is a graphical illustration of time with respect to frequency showing the typical group delay characteristics of the audio path in the filter-combiner herein. 
   FCC Mask Compliance 
   The FCC rules state, in part, “in the first 500 kHz from the authorized channel edge, transmitter emissions must be attenuated no less than 47 dB below the average transmitted power”. 
   Reference is now made to  FIG. 24  which illustrates a typical DTV transmitter spectral spread known in the prior art. The total integrated power between the vertical dotted lines  600  and  602  is the reference power for determining if the −47 dB requirement is met in the adjacent 0.5 MHz zones. 
     FIG. 25  illustrates the 0.5 MHz zone  604  of the typical DTV transmitter spectral spread illustration of the prior art. It is to be noted that spectral spread illustration of  FIG. 25  is without the employment of a sharp tuned filter, such as filter  100  described herein with reference herein to  FIG. 4  as well as with reference to the other versions illustrated in  FIGS. 7 ,  8 ,  11 ,  13  and  14 . To the contrary, the illustration in  FIG. 25  is representative of the prior art and it is clear that this does not meet the FCC −47 dB requirement. 
   Attention is now directed to the graphical illustration in  FIG. 26  of amplitude with respect to frequency showing the spectral spread when the sharp tuned filter  100  described herein is employed. At the location of the FCC 0.5 MHz adjacent zone  604 , the power is −51 dB below the in-band power level  610 . Clearly, this shows FCC mask compliance as more than the −47 dB level has been met noting specifically location  612  in the illustration of  FIG. 26 . 
   Moreover, it is noted from  FIG. 26  that the filter herein provides significant reduction of out of band spectrum components. 
   Power Enhancement 
   Reference is now made to  FIG. 27  which illustrates at curve  614 , the IMD (intermodulation distortion) sideband suppression characteristic without employing the sharply tuned filter herein. Curve  616  illustrates the sideband suppression characteristic with the sharp tuned filter herein when an IOT transmitter is operating at about 30% above normal power readings (note that the in-band power level  610 ′ in  FIG. 27  is about 30% greater than in-band power level  610  in  FIG. 26 ). 
   Reference is now made to  FIG. 28  which is similar to  FIG. 4 , but includes an adaptive feedback path  700  and a DTV exciter  702 . The exciter  702  includes a pre-corrector circuit  704  that pre-corrects the information signal supplied to antenna or load  106  to correct for any distortion caused by the filter  100 . The pre-correction may be for both non-linear and linear distortions. Preferably the pre-correction is for at least any linear distortion introduced by the filter  100 . As shown, the pre-correction may be adaptive with a feedback path. The pre-correction may also be non-adaptive, without the feedback path. The pre-correction should compensate for any linear distortion, such as any ripple in the in-band power level  120  in  FIG. 5  caused by filter  100  during such higher power operation noted in the waveform of  FIG. 27 . 
   Summation 
   
       
       1. A sharp tuned bandpass filter (STF) is a key element in a TV combining system for adjacent channel allocations where special techniques are required to combine very closely spaced channels. Current industry techniques use a set of notch diplexing cavities on the NTSC aural carrier in a constant impedance configuration to combine adjacent, upper or lower, DTV channels with an NTSC channel. The filter-combiner described herein at  FIG. 8  uses a set of sharp tuned filters to do a very effective job of combining the NTSC and DTV signals with significantly enhanced performance on the NTSC. 
       2. The NTSC aural performance is improved with nearly double the bandwidth of the notch cavity diplexing approach. This avoids sideband cutting of the “PRO Channel” and group delay and amplitude distortion of the SAP and BTSC Stereo subchannels. The notch diplexing approach introduces an “S” curve in the group delay response which makes it very difficult to correct. In addition, the amplitude response is sloped off in an asymmetrical manner further complicating the overall correction of the aural carrier. The sharp tuned filter herein eliminates “S” curve response problems with a soft group delay response that in many cases doesn&#39;t need correction or very little of it. 
       3. Video performance is enhanced with very little amplitude distortion and only mild group delay distortion which can be easily corrected. This is unlike the notch diplexing system that requires amplitude and delay correction circuits to restore the severe amplitude roll off of the 4.2 MHz burst flag due to the notch cavities. The sharp tuned filter system needs only modest group delay correction to restore the 12.5T pulse performance that can be easily accomplished with existing analog circuits. 
       4. The DTV to NTSC isolation characteristic is significantly degraded in the notch cavity system which causes high frequency bleed through of DTV components into the NTSC channel. The sharp tuned filter system described herein does not have this problem and provides a flat, uniform isolation value of about −35 dB over the video band. 
       5. The specified response and attenuation values shown herein provide the necessary symmetry in the DTV channel for elementary adaptive group delay and amplitude response correcting circuits to achieve automatic adaptive correction without losing control. The symmetry issue as specified is an important issue to reach the standard accepted error vector magnitude for DTV transmission. 
       6. The sharp tuned filter approach can combine adjacent channel DTV signals. 
       7. The attenuated point at +/−0.25 MHz off the upper and lower edge of the filter provides additional attenuation to meet the FCC out of band mask when the transmitter power is increased to a point where the spectral spread shoulder level can be as great as −30 dB. Full mask compliance can therefore be obtained when the DTV transmitter is operated at higher than normal power levels for enhanced efficient operation. The sharp tuned filter herein provides this characteristic. The benefit of higher power operation while maintaining FCC mask compliancy is a significant advantage. 
       8. The sharp tuned filter system with the attenuation specification shown herein exceeds the current FCC mask requirements and provides further sideband reduction for noticeably improved interference free operation on adjacent NTSC channels. 
       9. The system design concept of using a sharp tuned filter system for the unique application of adjacent upper or lower DTV channel combining with an NTSC station, only recently allocated by the FCC, with the performance as stated above. 
       10. A sharp tuned filter system causes very little group delay and amplitude errors over the video pass band and in particular, has a very smooth response curve over the aural pass band to maintain good BTSC performance and low synchronous AM. 
       11. The filter-combiner herein can combine adjacent DTV channels. An adaptive system can be connected to the system for adaptive correction on each independent DTV path. A special filter in the adaptive sample back line can be used in the exciter to filter out the adjacent DTV interference for excellent dual DTV transmission. 
       12. The filter herein, however, does restrict the 8VSB bandwidth slightly which is a benefit for reduced adjacent channel interference but does not cause any reduced performance on DTV receivers. There is no measurable change in DTV receiver threshold levels, hence no coverage loss. Transmitter EVM performance remains excellent. 
     
  
   From the above description, those skilled in the art will perceive improvements, changes and modifications. Such improvements, changes and modifications within the skill of the art are intended to be covered by the appended claims.