Patent Publication Number: US-2023137346-A1

Title: Adaptive rectification for preventing current inversion in motor driving

Description:
BACKGROUND 
     Technical Field 
     This application is directed to adaptive rectification for preventing current inversion in motor winding and, in particular, adaptive rectification that reduces power dissipation in the motor. 
     Description of the Related Art 
     Current inversion in motor windings can occur due to a back electromotive force voltage induced on the windings by rotation of the motor. Current is susceptible to inversion when the motor is operated with a small load and the average current passing through the windings is correspondingly small. Further, the current is susceptible to inversion when the motor is operated at high speed and the back electromotive force is correspondingly high. In addition, when a motor has a relatively small inductance, a high current ripple may result in current inversion. 
     BRIEF SUMMARY 
     A motor is operated by energizing windings of the motor and sourcing current in a first direction to a first end of the windings and sinking current from a second end of the windings. Energizing the windings occurs during a first time period that corresponds to an on time of a duty cycle of a control signal (pulse width modulated (PWM) signal) used to dictate operation of the motor. During a remainder of the duty cycle (an off time), active energizing of the windings ends. Instead, current of the windings is recirculated in the first direction through the windings. 
     Current is susceptible to reversal of direction to a second, opposite, direction during the recirculation period. If the current reverses direction, the motor operates less efficiently. Thus, it is advantageous to prevent current from recirculating after the current reverses direction to the second direction. In addition, it is advantageous to allow current to recirculate through a bi-directional conductive path of a transistor (that is in the conductive state) rather than a body diode or a free-wheeling diode of the transistor (that is in the non-conductive state). 
     During the recirculation period, current is initially permitted to recirculate in the first direction through the conductive path of the transistor, and the direction of current is monitored. If the current reverses direction to the second direction, the transistor is switch off to prevent the current from passing through the transistor in the reverse direction. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
         FIG.  1    shows a system including a motor and a control stage. 
         FIG.  2    shows an electrical modeling of the phases of the motor. 
         FIG.  3 A  shows an example of control of a half-bridge stage. 
         FIG.  3 B  shows an example of control of a half-bridge stage. 
         FIGS.  4 A and  4 B  show power dissipation by the motor, power dissipation by a high side transistor and power dissipation by a low side transistor in relation to current passing through windings when the half-bridge stage is operated according to the adaptive rectification technique. 
         FIGS.  5 A and  5 B  show power dissipation by the motor, power dissipation by the high side transistor and power dissipation by the low side transistor in relation to the current passing through the windings when the half-bridge stage is operated according to synchronous rectification. 
         FIGS.  6 A and  6 B  shows power dissipation by the motor, power dissipation by the high side transistor and power dissipation by the low side transistor in relation to the current passing through the windings when the half-bridge stage is operated according to quasi-synchronous rectification. 
         FIG.  7    shows an example of control of the half-bridge stage when current is recirculated through the high side transistors. 
         FIG.  8 A  shows the decay mode selection stage according to an embodiment. 
         FIG.  8 B  shows the decay mode selection stage according to an embodiment. 
     
    
    
     DETAILED DESCRIPTION 
       FIG.  1    shows a system  100  including a motor  102  and a control stage  104 . The control stage  104  includes a sequencing stage  106 , a driving stage  108 , a decay mode selection stage  110  and a half-bridge stage  112 . The motor  102  may be a multi-phase (e.g., three-phase) brushless direct current (DC) motor (BLDC). Thus, the motor  102  may include a plurality of phases that are denoted herein as U, V, W. 
     The half-bridge stage  112  includes a plurality of half-bridges  114   a ,  114   b ,  114   c  respectively corresponding to the plurality of phases of the motor  102 . Each half-bridge  114   a ,  114   b ,  114   c  includes a respective high side transistor  116   a ,  116   b ,  116   c  and a respective low side transistor  118   a ,  118   b ,  118   c . The high side transistors  116   a ,  116   b ,  116   c  each have a first conduction terminal coupled to a voltage supply node  120  and a second conduction terminal coupled to a respective half-bridge node  122   a ,  122   b ,  122   c . The low side transistors  118   a ,  118   b ,  118   c  each have a first conduction terminal coupled to the respective half-bridge node  122   a ,  122   b ,  122   c  and a second conduction terminal coupled to ground  124 . As shown in  FIG.  1   , a sense resistor  126  may be coupled between the second conduction terminals of the low side transistor  118   a ,  118   b ,  118   c  and ground  124  for generating a current sense voltage representative of current passing therethrough. The high side transistors  116   a ,  116   b ,  116   c  and the low side transistor  118   a ,  118   b ,  118   c  each have a control terminal coupled to the driving stage  108 . Furthermore, the transistors  116   a ,  116   b ,  116   c ,  118   a ,  118   b ,  118   c  each have a free-wheeling diode (body diode) in parallel therewith. 
     The sequencing stage  106  has outputs coupled to inputs of the driving stage  108 , respectively, and inputs of the decay mode selection stage  110 , respectively. The driving stage  108  has outputs coupled to the control terminals of the high side transistors  116   a ,  116   b ,  116   c  and the low side transistor  118   a ,  118   b ,  118   c , respectively. The decay mode selection stage has inputs coupled to the half-bridge nodes  122   a ,  122   b ,  122   c , respectively. The decay mode selection stage  110  has an output coupled to an input of the driving stage  108 . The sequencing stage  106 , driving stage  108  and decay mode selection stage  110  may be part of one or more controllers of the motor  102 . 
     The sequencing stage  106  generates and outputs one or more enable signals enabling commutation of a half-bridge corresponding to a phase of the motor  102 . The sequencing stage  106  also outputs a pulse width modulation (PWM) signal representing the PWM cycle of the commutation. The sequencing stage  106  may also output, to the driving stage  108 , a signal indicating a desired direction of current in the motor  102 . The driving stage  104  receives the one or more enable signals and the PWM signal and controls the high side transistors  116   a ,  116   b ,  116   c  and the low side transistor  118   a ,  118   b ,  118   c  based on the one or more enable signals and the PWM signal. Controlling the high side transistors  116   a ,  116   b ,  116   c  and the low side transistor  118   a ,  118   b ,  118   c  includes sending driving signals (HSU, HSV, HSW, LSU, LSV, LSW) to the respective control terminals of the high side transistors  116   a ,  116   b ,  116   c  and the low side transistor  118   a ,  118   b ,  118   c  to put the transistors in conductive or non-conductive state. 
     If the one or more enable signals enable the first half-bridge  114   a  (phase U), the driving stage  104  controls the first half-bridge  114   a  to commutate according to the PWM signal conditional upon the current passing through the phase. The driving stage  104  puts the low side transistor  118   b  of the second half-bridge  114   b  (phase V) in the conductive state (or alternatively puts the high side transistor  116   b  of the second half-bridge  114   b  in the conductive state if the high side transistors  116   a ,  116   b ,  116   c  are used for current recirculation). If the current reverses direction, the driving stage  104  controls the first half-bridge  114   a  and the second half-bridge  114   b  as described herein according to the adaptive rectification. The driving stage  104  puts the third half-bridge  114   c  (phase W) in a high impedance state with both the high and low side transistors  116   c ,  118   c  turned off. The driving stage  104  similarly alternates between different pairs of half-bridges for current passage depending on the half-bridge indicated as enabled by the one or more enable signals. 
     The decay mode selection stage  110  receives a plurality of signals. A signal of the plurality of signals is representative of a voltage of a respective half-bridge node  122   a ,  122   b ,  122   c . The decay mode selection stage  110  compares the voltages of two half-bridge nodes  122   a ,  122   b ,  122   c  used for current circulation. A result of the comparison is representative of a direction of current in a phase of the motor  102 . Based on the comparison, the decay mode selection stage  110  outputs a control signal to the driving state  108 . The control signal indicates to the driving state  108  the direction of current flow in the phase or whether to operate the half-bridge stage  112  in synchronous (first mode) or quasi-synchronous (second mode) rectification. As described herein, the driving state  108  operates the transistors  116   a ,  116   b ,  116   c ,  118   a ,  118   b ,  118   c  based on the control signal. The driving state  108  operates the transistors to block current recirculation when the current reverses direction. 
     A plurality of windings of the motor  102  may be positioned on a stator (not shown) of the motor  102 . The motor  102  may be a synchronous motor, and a rotation speed of the motor may be synchronized with a frequency of current in the windings. Rotation is obtained in response to the magnetic field generated by the current passing through the plurality of windings. 
     The plurality of windings each have respective first and second ends. The plurality of windings may be coupled in a star configuration in the motor, whereby the respective first ends of the plurality of windings are coupled to each other. The second ends of the plurality of windings are respectively coupled to the half-bridge nodes  122   a ,  122   b ,  122   c . Displacement between the magnetic field generated by the current passing through the plurality of windings and a magnetic field generated by the rotor&#39;s permanent magnets exerts a torque that produces the motor rotation. 
       FIG.  2    shows an electrical modeling of the phases of the motor  102 . The winding of each phase U, V, W of the motor  102  may be represented as an inductance  128   a ,  128   b ,  128   c  (denoted ‘Lm’), a resistance  130   a ,  130   b ,  130   c  (denoted ‘Rm’) and a voltage source  132   a ,  132   b ,  132   c  (denoted ‘V BEMF (t)’) that are serially coupled. As described herein, a first end of a winding is coupled to a star configuration node  134  and a second end of the winding serves as a motor lead and is coupled to a respective half-bridge node  122   a ,  122   b ,  122   c . The voltage source  132   a ,  132   b ,  132   c  represents a back electromotive force voltage induced on the windings by motor rotation. A level of the back electromotive force voltage is proportional to a rotation speed of the rotor. When an angular displacement between rotor and stator magnetic fields is around 90°, the back electromotive force voltage opposes the current passing through the winding. 
     Torque is positively correlated with an average of the current passing through the windings (I phase,avg ). In addition, torque depends on the angular displacement between the rotor and stator magnetic fields. Generally, torque is maximized when the angular displacement is 90°. Torque reaches a minimum when the magnetic fields are aligned and the angular displacement is 0°. The torque may be represented as: 
         Tq∝I   phase,avg   ×f (θ).  Equation (1)
 
     where f(θ) is a function relating rotor and stator magnetic field displacement to torque. 
     The control stage  104  may drive the motor  102  according to a six-step sequence technique in which one pair of windings is energized at a time according to a sequence of six possible current directions, each of which corresponding to a stator magnetic field vector. The control stage  104  may commutate current to different pairs of windings based on a sensor reading or sensorless techniques, such as back electromotive force zero-crossing detection. Commutating the current allows for keeping the angular displacement between the two magnetic fields in a desired optimal range. The control stage  104  may select a subsequent pair to energize based on a desired motor direction. 
     The driving stage  108  may operate the transistors  116   a ,  116   b ,  116   c ,  118   a ,  118   b ,  118   c  between conductive and non-conductive states stage based on the one or more PWM signals according to the six-step sequence technique. Further, the driving stage  108  may adjust the average current (I phase,avg ), directly or indirectly, to regulate the torque. 
       FIG.  3 A  shows an example of control of the half-bridge stage  112 . First and second half-bridges  114   a ,  114   b  corresponding respective sides (a first side (side  1 ) and a second side (side  2 )) are shown in  FIG.  3 A . It is noted that although the first and second half-bridges  114   a ,  114   b  corresponding to phases U and V, respectively, are shown in  FIG.  3 A , the control described herein may be used for any other pair of half-bridges, phases or sides used to operate the motor  102 . 
     The decay mode selection stage  110  includes a first comparator  136   a . The first comparator  136   a  has a first input (for example, a non-inverting input) coupled to the first half-bridge node  122   a  and a second input (for example, an inverting input) coupled to the second half-bridge node  122   b . The first comparator  136   a  also has an output. A pair of windings  138   a  corresponding to the windings of the U and V phases is coupled between the half-bridge nodes  122   a ,  122   b . The pair of windings  138   a  are modeled as an inductance, resistance and voltage source that are serially coupled. The inductance, resistance and voltage source of the pair of windings  138   a  are aggregates of the inductances  128   a ,  128   b , resistances  130   a ,  130   b  and voltage sources  132   a ,  132   b , respectively. 
     While the first and second half-bridges  114   a ,  114   b  are operated during a PWM cycle, the third half-bridge  114   c  is in a high impedance state. During the high impedance state, the third high side transistor  116   c  and the third low side transistor  118   c  of the third half-bridge  114   c  are non-conductive (switched off). 
     During an on time of the PWM cycle, the driving stage  108  puts the first high side transistor  116   a  in a conductive state and the first low side transistor  118   a  in a non-conductive state. In addition, the driving stage  108  puts the second high side transistor  116   b  in a non-conductive state and the second low side transistor  118   a  in a conductive state. During the on time, the driving stage  108  charges the pair of windings  138   a  by applying a supply voltage of the voltage supply node  120  to the windings  138   a . Current flows through the windings  138   a  from the first half-bridge node  122   a  to the second half-bridge node  122   b . The first half bridge  114   a  sources current and the second half bridge  114   b  sinks current. 
     When the on time of the PWM cycle ends and the off time of the PWM cycle begins, the driving stage  108  switches operation from current charging to current recirculation using synchronous rectification. At the start of the off time, the driving stage  108  transitions the first high side transistor  116   a  to the non-conductive state and the first low side transistor  118   a  to the conductive state. The driving stage  108  maintains the second half-bridge  114   b  in the same state as the on time. Thus, the driving stage  108  shorts the second ends of the windings  138   a . The shorting allows the windings  138   a  to discharge. 
     During each of the commutations described above, the driving stage  108  may introduce an intermediate period, generally named dead-time, where both the high side transistor  116   a  and the low side transistor  118   a  of the first half-bridge  114   a  are in the non-conductive state. 
     The current passing through the windings  138   a  may reverse direction. Direction reversal may occur due to fact that the current is not constant in magnitude. The current is susceptible to ripples resulting from characteristics of the motor, the supply voltage and the back electromotive force. In particular, when the motor  102  is operated at a relatively low torque and using a corresponding low average current (I phase,avg ), current inversion is more likely to occur. For example, a low average current may be below one ampere (A). 
     If the current reverses direction, the driving stage  108  does not use synchronous rectification to operate the half-bridge stage  112  during the entirety of the off time. The driving stage  108  ends synchronous rectification (first mode of rectification) and operates the half-bridge stage  112  using quasi-synchronous rectification (second mode of rectification) in response to current direction reversal. 
     During synchronous rectification, the decay mode selection stage  110  compares the voltages at the ends of the windings  138   a  to identify a direction of the current passing through the windings  138   a . As shown in  FIG.  3 A , the first comparator  136   a  of the decay mode selection stage  110  receives a first voltage of the first half-bridge node  122   a  and a second voltage of the first half-bridge node  122   b . The first voltage may be representative of a voltage drop (VDS1) across the first low side transistor  118   a , and the second voltage may be representative of a voltage drop (VDS2) across the second low side transistor  118   b . The first voltage may be a product of the current passing through the windings  138   a  and a total resistance between the conductive terminals of the first low side transistor  118   a  when the first low side transistor  118   a  is conductive (also referred to as a drain-source on resistance (R DS(on) ). The second voltage may be a product of the current passing through the windings  138   a  and a drain-source on resistance (R DS(on) ) of the second low side transistor  118   a.    
     The second conduction terminals of the low side transistors  118   a ,  118   b  are coupled to each other. Thus, if the first voltage is less than the second voltage, then the current has not reversed direction. The current flows from the second low side transistor  118   b  to the first low side transistor  118   a  through their common coupling and from the first low side transistor  118   a  to the second low side transistor  118   b  through the windings  138   a . If the current has not reversed direction, the decay mode selection stage  110  keeps operating the half-bridge stage  112  using synchronous rectification. The decay mode selection stage  110  continues comparing the first and second voltages during the off time. 
     Conversely, if the first voltage is greater than the second voltage, then the current has reversed direction. The current flows from the first low side transistor  118   a  to the second low side transistor  118   b  through their common coupling and from the second low side transistor  118   b  to the first low side transistor  118   a  through the windings  138   a . If the first and second voltages are the same, then current does not flow through the windings  138   a  (or the current level is OA). 
     If the current reverses direction, the decay mode selection stage  110  responds by dynamically causing operation to transition to quasi-synchronous rectification. The decay mode selection stage  110  outputs the control signal commanding the driving stage  108  to switch off the first low side transistor  118   a . The first low side transistor  118   a  becomes non-conductive and prevents the current from flowing in the reverse direction. 
     Alternatively, the driving stage  108  switches off the second low side transistor  118   b . The second low side transistor  118   b  becomes non-conductive and prevents the current from flowing in the reverse direction as shown in the example of control shown in  FIG.  3 B . 
     The decay mode selection stage  110  may compare the first and second voltages and output the control signal during an off time of a PWM cycle of the PWM signal. The decay mode selection stage  110  may refrain from comparing the first and second voltages and outputting the control signal during an on time of the PWM cycle. Further, the decay mode selection stage  110  may cause operation to transition from quasi-synchronous rectification to synchronous rectification in response to detecting that direction reversal has ceased during the off time of the PWM cycle. 
     Operating the half-bridge stage  112  using adaptive rectification as described herein is advantageous over operating the half-bridge stage  112  exclusively using synchronous rectification or exclusively using quasi-synchronous rectification during the off time. Operating the half-bridge stage  112  using adaptive rectification results in less power dissipation as compared to operating exclusively using quasi-synchronous rectification. 
     Further, operating the half-bridge stage  112  prevents current inversion observed in synchronous rectification. 
     Power dissipation by the windings  138   a  is positively correlated with the root mean square (RMS) of the current passing through the windings (I phase,RMS ). The power dissipation is represented as: 
         P   d   ∝I   phase,RMS   Equation (2)
 
     Current inversion reduces the average of the current passing through the windings (I phase,avg ) resulting in lower torque. However, current inversion positively contributes to the RMS of the current and results in increasing power dissipation. Preventing current inversion reduces the mismatch between the average and the RMS of the current and improves the efficiency of the motor  102 . 
     Furthermore, the power dissipated in response to the passage of current through the free-wheeling diode of the first low side transistor  118   a  (when the first low side transistor  118   a  is conductive) is higher than the power dissipated as a result of the current passage through the conductive path of the first low side transistor  118   a  (when the first low side transistor  118   a  is non-conductive). 
     In exclusive synchronous rectification, the first low side transistor  118   a  is conductive during the off time. Recirculation current passes through the first low side transistor  118   a . Thus, less power is dissipated as a result of the passage of the recirculated current in the first low side transistor  118   a  than the alternative use of the free-wheeling diode to allow current passage. However, when the current reverses direction, the first low side transistor  118   a  is in a conductive state, and passage of the current in the reverse direction is permitted. 
     Alternatively, in exclusive quasi-synchronous rectification, the first low side transistor  118   a  is non-conductive during the off time. The current recirculates through the free-wheeling diode of the first low side transistor  118   a . The free-wheeling diode advantageously blocks the flow of current if the current is inverted and reverses direction. However, the free-wheeling diode dissipates more power than the first low side transistor  118   a  in the conductive state. Thus, less power is dissipated as a result of the passage of the recirculated current through the first low side transistor  118   a  than the alternative use of the free-wheeling diode. However, when the current reverses direction during synchronous rectification, the first low side transistor  118   a  is in a conductive state, and passage of the current in the reverse direction is permitted. 
       FIG.  4 A  shows power dissipation  402  by the motor  102 , power dissipation  404  by the high side transistor  116   a  and power dissipation  406  by the low side transistor  118   a  in relation to the current  408  passing through the windings  138   a  when the half-bridge stage  112  is operated according to the adaptive rectification technique described herein. The inductance (Lm)  128   a  is 100 microhenry (μH), the resistance (Rm)  130   a  is 0.1 Ohm (Ω), the voltage source (V BEMF (t))  132   a  is 15 volts (V) and the supply voltage is 24V. To facilitate a higher torque, the current  408  passing through the windings  138   a  is relatively high. The current  408  passing through the windings  138   a  is maintained at above 7.5 A. 
     Due to the elevated current  408 , the risk of current inversion is low. The power  402  dissipated by the motor  102  varies in relation to the current  408  passing through the windings  138   a . Further, the high side transistor  116   a  dissipates power  404  during the on time of the PWM cycle when the high side transistor  116   a  is conductive and ceases dissipating power  404  during the off time of the PWM cycle when the high side transistor  116   a  is non-conductive. Conversely, the low side transistor  118   a  dissipates power  406  during the off time of the PWM cycle when the low side transistor  118   a  is conductive and ceases dissipating power  406  during the on time of the PWM cycle when the low side transistor  118   a  is non-conductive. 
     Due to the absence of current inversion in  FIG.  4 A , the mean and RMS of the current  408  passing through the windings  138   a  are commensurate with each other. The low side transistor  118   a  does not dissipate power in an excessive manner. 
       FIG.  4 B  shows the power dissipation  402  by the motor  102 , the power dissipation  404  by the high side transistor  116   a  and the power dissipation  406  by the low side transistor  118   a  in relation to the current  408  passing through the windings  138   a  when the half-bridge stage  112  is operated according to the adaptive rectification technique described herein. The inductance (Lm)  128   a , resistance (Rm)  130   a , voltage source (V BEMF (t))  132   a  and supply voltage is 24V have the same values described with reference to  FIG.  4 A . The current  408  passing through the windings  138   a  is lower than the current  408  of  FIG.  4 A  (for example, due to the motor being operated at lower torque). 
     As shown in  FIG.  4 B , the current  408  is prevent from being inverted. The motor  102  dissipates power  402  when the current  408  is passing through the windings  138   a . Further, the low side transistor  118   a  exhibits a peak in power dissipation  406  when the low side transistor  118   a  is switched from on to off. The power dissipation  402 ,  404 ,  406  of the motor  102 , high side transistor  116   a  and low side transistor  118   a , respectively, are negligible when the current is OA (and is blocked from flowing through the windings  138   a  due to direction reversal). 
     During the operation of  FIGS.  4 A and  4 B  using the adaptive rectification technique described herein, the power dissipation  402  by the motor  102  is the predominant contributor to the total dissipated power. The power dissipation  404  by the high side transistor  116   a  and the power dissipation  406  by the low side transistor  118   a  are comparatively small in relation to the power dissipation  402  by the motor  102 . 
       FIG.  5 A  shows power dissipation  502  by the motor  102 , power dissipation  504  by the high side transistor  116   a  and power dissipation  506  by the low side transistor  118   a  in relation to the current  508  passing through the windings  138   a  when the half-bridge stage  112  is operated according to synchronous rectification. The values of the inductance (Lm)  128   a , resistance (Rm)  130   a , voltage source (V BEMF (t))  132   a  and the supply voltage are the same as described with reference to  FIG.  4 A . The current  508  passing through the windings  138   a  remains positive and is not blocked as a result of inversion. Accordingly, the power dissipation performance of synchronous rectification is the same as the power dissipation performance of the adaptive rectification technique shown in  FIG.  4 A . The power dissipation  502  by the motor  102 , the power dissipation  504  by the high side transistor  116   a  and the power dissipation  506  by the low side transistor  118   a  are the same as the power dissipation  402  by the motor  102 , the power dissipation  404  by the high side transistor  116   a  and the power dissipation  406  by the low side transistor  118   a , respectively, described with reference to  FIG.  4 A . 
       FIG.  5 B  shows the power dissipation  502  by the motor  102 , the power dissipation  504  by the high side transistor  116   a  and the power dissipation  506  by the low side transistor  118   a  in relation to the current  508  passing through the windings  138   a  when the half-bridge stage  112  is operated according to synchronous rectification. The inductance (Lm)  128   a , resistance (Rm)  130   a , voltage source (V BEMF (t))  132   a  and supply voltage is 24V have the same values described with reference to  FIG.  4 A . As shown in  FIG.  5 B , the current  508  has a lower value average value than the current  508  shown in  FIG.  5 A , and, thus, the current  508  is more susceptible to direction reversal. 
     As shown in  5 B, the current  508  reverses direction during operation of the motor  102 . The motor  102  dissipates power  502  when the current  508  is passing through the windings  138   a  irrespective of the direction of the current  508  resulting in increased power dissipation compared to operation using the adaptive rectification technique described herein. Further, the high side transistor  116   a  and the low side transistor  118   a  also dissipate more power. 
       FIG.  6 A  shows power dissipation  602  by the motor  102 , power dissipation  604  by the high side transistor  116   a  and power dissipation  606  by the low side transistor  118   a  in relation to the current  608  passing through the windings  138   a  when the half-bridge stage  112  is operated according to quasi-synchronous rectification. The values of the inductance (Lm)  128   a , resistance (Rm)  130   a , voltage source (V BEMF (t))  132   a  and the supply voltage are the same as described with reference to  FIG.  4 A . The current  608  passing through the windings  138   a  remains positive and is not blocked as a result of inversion. Accordingly, the power dissipation of the motor  102  and the high side transistor  116   a  are comparable to the power dissipation performance of the adaptive rectification technique shown in  FIG.  4 A . 
     However, the low side transistor  118   a  dissipates more power during the off time of the PWM duty cycle due to the fact that the low side transistor  118   a  is turned off and the current  608  passes through the body diode of the low side transistor  118   a . The body diode of the low side transistor  118   a  dissipates more power than the conductive path of the low side transistor  118   a.    
       FIG.  6 B  shows the power dissipation  602  by the motor  102 , the power dissipation  604  by the high side transistor  116   a  and the power dissipation  606  by the low side transistor  118   a  in relation to the current  608  passing through the windings  138   a  when the half-bridge stage  112  is operated according to quasi-synchronous rectification. The inductance (Lm)  128   a , resistance (Rm)  130   a , voltage source (V BEMF (t))  132   a  and supply voltage is 24V have the same values described with reference to  FIG.  4 A . 
     Similar to the current  608  in  FIG.  6 A , during the off time of the PWM cycle, the low side transistor  118   a  dissipates more power than in the adaptive rectification technique of  FIG.  4 B  due to the fact that the low side transistor  118   a  is turned off and the current  608  passes through the body diode of the low side transistor  118   a . As compared to  FIG.  5 B , the quasi-synchronous rectification blocks the current reversal. 
     The adaptive rectification technique described herein avoids current inversion. The adaptive rectification technique maximizes a ratio between the average and the RMS of the current passing through the windings of a motor to minimize power dissipation. The current is recirculated into switches rather than free-wheeling diodes thereof to reduce power dissipation. 
     In an embodiment, the driving stage  108  or the decay mode selection stage  110  may use filtering and/or blanking to suppress or disable transitions between quasi-synchronous rectification and synchronous rectification that occur within a defined period of time. The use of filtering and/or blanking is advantageous in that it prevents frequent switching between modes of operation. 
     In an embodiment, the driving stage  108  may transition from synchronous to quasi-synchronous operation at most once during an off time of a PWM cycle. Thus, when the control signal indicates that operation is to be switched from synchronous to quasi-synchronous operation during an off time, the driving stage  108  switches to quasi-synchronous operation and keeps operating the half-bridge stage  112  quasi-synchronously until the end of the off time. 
     It is noted that lower power dissipation allows for increasing the density of the control stage  104  (for example, on a printed circuit board (PCB). Further, lower power dissipation allows for using fewer heat sinks, using fewer ventilation devices, forgoing the use of heat sinks and forgoing the use of ventilation devices to cool the control stage  104 . In addition, reducing the power dissipation reduces manufacturing costs as it permits usage of transistors with larger drain-source on resistance (R DS(on) ) in the half-bridge stage  112  that are more economical than counterparts with a lower drain-source on resistance (R DS(on) ). 
     Although embodiments are described herein in which the current passing through the windings of the motor  102  recirculates through the low side transistors  118   a ,  118   b ,  118   c , a different convention may be adopted. For example, the current may be recirculated through the high side transistors  116   a ,  116   b ,  116   c.    
       FIG.  7    shows an example of control of the half-bridge stage  112  when current is recirculated through the high side transistors  116   a ,  116   b ,  116   c . Similar elements of  FIG.  7    as those described with reference to  FIG.  3 A  have the same reference numerals. In  FIG.  7   , the first and second low side transistors  118   a ,  118   b  are non-conductive and the first and second high side transistors  116   a ,  116   b  are commonly coupled at the voltage supply node  120  (having the supply voltage). The first voltage may represent a difference between the supply voltage and a voltage drop (VDS1) across the first high side transistor  116   a , and the second voltage may be representative of a difference between the supply voltage and a voltage drop (VDS2) across the second high side transistor  116   b.    
     When the first voltage is lower than the second voltage, the current flows from the first high side transistor  116   a  into the windings  138   a  and from the windings  138   a  to the second high side transistor  116   b . Conversely, when the first voltage is higher than the second voltage, the current flows from the second high side transistor  116   b  into the windings  138   a  and from the windings  138   a  to the first high side transistor  116   a . If the desired current direction is counterclockwise in the half-bridge stage  112  of  FIG.  7   , then the current reverses direction when the first voltage is higher than the second voltage. Thus, during off time operation, the decay mode selection stage  110  switches operation from synchronous rectification to quasi-synchronous rectification in response detecting that the first voltage has become higher than the second voltage. It is noted, various operations described herein in relation to current recirculation through the low side transistors  118   a ,  118   b ,  118   c  may be changed to accommodate current recirculation through the high side transistors  116   a ,  116   b ,  116   c.    
       FIG.  8 A  shows a decay mode selection stage  110  according to an embodiment. The decay mode selection stage  110  is shown coupled to the half-bridge stage  112 . In addition to the first comparator  136   a , the decay mode selection stage  110  includes second and third comparators  136   b ,  136   c . The second comparator  136   b  has a first input (for example, a non-inverting input) coupled to the second half-bridge node  122   b  and a second input (for example, an inverting input) coupled to the third half-bridge node  122   c . The second comparator  136   b  also has an output. The third comparator  136   c  has a first input (for example, a non-inverting input) coupled to the first half-bridge node  122   a  and a second input (for example, an inverting input) coupled to the third half-bridge node  122   c . The third comparator  136   c  also has an output. 
     As described in relation to the first comparator  136   a  herein, the second comparator  136   b  identifies current reversal when the second and third phases are energized (for example, per the six-step sequence technique). The third comparator  136   c  identifies current reversal when the first and third phases are energized. The comparison results output by the comparators  136   a ,  136   b ,  136   c  are used by the decay mode selection stage  110  to output the control signal to the driving stage  108 . 
     In place of utilizing multiple comparators respectively corresponding to different energized phase pairs, the decay mode selection stage  110  may include switches that selectively couple comparator inputs to different phases. 
       FIG.  8 B  shows a decay mode selection stage  110  according to an embodiment. The decay mode selection stage  110  includes a comparator  140  and first and second switches  142   a ,  142   b . The comparator  140  has first and second inputs and an output. The first switch  142   a  has a first terminal coupled to the first input of the comparator  140 , and the second switch  142   b  has a first terminal coupled to the second input of the comparator  140 . The first switch  142   a  has second and third terminals respectively coupled to the first and second half-bridge nodes  122   a ,  122   b . The second switch  142   b  has second and third terminals respectively coupled to the second and third half-bridge nodes  122   b ,  122   c.    
     The first and second switches  142   a ,  142   b  operate to respectively couple the first and second inputs of the comparator  140  to two different half-bridge nodes  122   a ,  122   b ,  122   c  used to recirculate current through the windings  138   a . The comparison result output by the comparator  140  are used by the decay mode selection stage  110  to output the control signal to the driving stage  108 . 
     The various embodiments described above can be combined to provide further embodiments. These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure.