Patent Publication Number: US-9844099-B2

Title: Induction heating apparatus

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of Korean Application No. 10-2014-0004805, filed Jan. 14, 2014, in the Korean Intellectual Property Office, the disclosure of which is incorporated herein by reference. 
     BACKGROUND 
     1. Field 
     Embodiments of the present disclosure relate to an induction heating apparatus with an inverter. 
     2. Description of the Related Art 
     In comparison with conventional gas burners and oil burners having been used as heaters, the induction heating apparatus has short heating time which leads to less possibility for deformation by heat, and causes self-heating which leads to high efficiency, thereby enabling an entire system to be compact in size and light in weight. Modern high-frequency induction heating apparatuses have a trend to be implemented digitally taking into account protection of switching devices and improved reliability and accuracy. Recent development of power semiconductors, such as Insulated Gate Bipolar mode Transistors (IGBTs), Integrated Gate-Commutated Thyristors (IGCTs), Silicon Controlled Rectifiers (SCRs), Metal-Oxide Semiconductor Field Effect Transistors (MOS-FETs) has enabled resonant converters and inverters in need of high frequency switching to be designed. 
     The induction heating apparatus is largely divided into two different types. First, there is an induction heating apparatus with a voltage source inverter. The voltage source inverter type induction heating apparatus regulates an output by changing a switching frequency and performs switching at a higher frequency than a desired resonance frequency in a way to draw reactive power from within a resonance circuit. This provides an advantage of decreased reactive power due to the phase delay of a current with respect to a supply voltage and attainment of zero-voltage switching in an automatic turning on phase by using load resonance and switching the load at a frequency higher than the resonance frequency. 
     However, power loss increases as the switching frequency becomes higher. Accordingly, to reduce the power loss, a circuit, instead of an auxiliary circuit is required or a switching scheme, instead of the zero-voltage switching scheme, for reducing stress on the power semiconductor devices is required. 
     In contrast, the second type induction heating apparatus uses a high-frequency half-bridge inverter that has advantages of attaining the zero-voltage switching and having low voltage power semiconductors, by nature of the half-bridge scheme, thereby being easily expandable and thus having been widely used in induction heating systems. With the high-frequency half-bridge type inverter, fast heating may be achieved with high reliability, easy control over outputs and temperature, and less pollutants resulting from the heating procedure, thereby securing better heating efficiency, less heat loss and more clean surroundings. 
     Such an induction heating apparatus of a related art may include an RLC load block having e.g., inductance L from a coil for induction heating, resistance R of an object to be heated, capacitance C of a power factor compensative condenser, and main technologies of the heating induction apparatus are to find a resonance frequency to make impedance and admittance of the RLC load block minimum and thus to supply power to the load at an optimum power efficiency. In other words, it is an important factor that a resonance state is maintained even if the object to be heated (e.g., a pot, a vessel, etc.) has been moved due to a manipulation by a user at the beginning of driving the induction heating apparatus or during heating. 
     To do this, the induction heating apparatus of the related art measures RLC of a fixed load to select a suitable resonance frequency and performs non-changeable switching for the inverter. That is, since the switching signal sent to the inverter has a fixed resonance frequency, it is difficult to actively react to a situation changed with a change in load state and thus heating efficiency may degrade significantly. In other words, if there is a change in the RCL value due to various factors, resonance frequency deviation may occur and thus induction heating may not work at a maximum efficiency. 
     In addition, even a change in electrical characteristics due to the load in the induction heating apparatus may require a phase difference between voltage and current applied to a coil to be increased for robust current control, and in this case a switch turn-off current may increase, thereby increasing switching loss. 
     SUMMARY 
     Additional aspects and/or advantages will be set forth in part in the description which follows and, in part, will be apparent from the description, or may be learned by practice of the invention. 
     To address the problem, an objective of embodiments of the present disclosure is to implement optimum switching operations so as to minimize a phase difference between voltage and current applied to a coil and reduce switching losses. 
     In accordance with an aspect of the present disclosure, an induction heating apparatus is provided. The induction heating apparatus includes a coil; an inverter unit configured to have a switching device turned on and off to supply power to the coil; a controller configured to generate a threshold current based on information about a coil current flowing in the coil and an input voltage applied to the coil, and to generate a clock signal by comparing the coil current with the threshold current; and a switch driver configured to generate a switch driving signal to turn on or off the switching device of the inverter unit by dividing a frequency of the clock signal. A resonance point tracking controller and a switch driver may be implemented by a micom. 
     High level intervals of the clock signal may be made in intervals where the coil current is less than the threshold current. 
     The controller may include a comparator, wherein the comparator has magnitude information of the coil current at its inverting input and has the threshold current at its non-inverting input. 
     The threshold current may be determined to be proportional to a peak value of the coil current divide by the input voltage. 
     The threshold current may be determined to have a magnitude to keep turn-off current of the switching device minimum. 
     The switch driver may include a D flip-flop having the clock signal input to its clock input and a negative logic output connected to its data input; and a logic gate configured to generate a switch driving signal to drive the switching device of the inverter unit using a positive logic output and the negative logic output of the D flip-flop. 
     The logic gate may include an AND gate for generating the switch driving signal by dividing a frequency of the positive logic output of the D flip-flop approximately by two. 
     The inverter unit may comprise a half-bridge inverter. 
     The controller and the switch driver may constitute a half-bridge inverter implemented by a micom. 
     In accordance with another aspect of the present disclosure, an induction heating apparatus is provided. The induction heating apparatus includes a coil; an inverter unit configured to have a switching device turned on and off to supply power to the coil; a first controller configured to generate a first threshold current based on information about a coil current flowing in the coil and an input voltage applied to the coil, and to generate a clock signal by comparing the coil current with the first threshold current; a switch driver configured to generate a switch driving signal to turn on or off the switching device of the inverter unit by dividing a frequency of the clock signal; and a second controller configured to generate an enable signal for selectively restricting input of the clock signal to the switch driver based on output information set by a user, when the clock signal output from the first controller is input to the switch driver. 
     High level intervals of the clock signal may be made in intervals where the coil current is less than the first threshold current. 
     The first controller may include a first comparator, wherein the first comparator has magnitude information of the coil current at its inverting input and has the first threshold current at its non-inverting input. 
     The first threshold current may be determined to be proportional to a peak value of the coil current divide by the input voltage. 
     The first threshold current may be determined to have a magnitude to keep turn-off current of the switching device minimum. 
     The switch driver may include a D flip-flop having the clock signal input to its clock input and a negative logic output connected to its data input; and a logic gate configured to generate a switch driving signal to drive the switching device of the inverter unit using a positive logic output and the negative logic output of the D flip-flop. 
     The logic gate may include an AND gate for generating the switch driving signal by dividing a frequency of the positive logic output of the D flip-flop approximately by two. 
     The inverter unit may comprise a half-bridge inverter. 
     The second controller may include a second comparator, and wherein the second comparator has a Proportional Integral (PI) control value resulting from PI control over a difference between an absolute value of the coil current and a second threshold current at its non-inverting input, and has a sawtooth wave signal at its non-inverting input. 
     The second threshold current may represent a target output of the induction heating device set by a user. 
     The second controller may generate a pulse signal as the enable signal having a high level value for intervals where the PI control value is greater than the sawtooth wave signal and a low level value for intervals where the PI control value is equal to or less than the sawtooth wave signal. 
     A frequency of the clock signal may be divided in the switch driver in the high level intervals of the enable signal. 
     The first controller, the switch driver, and the second controller may constitute a half-bridge inverter implemented by a micom. 
     In accordance with a further aspect of the present disclosure, an induction heating apparatus includes a coil; an inverter unit configured to have a switching device turned on and off to supply power to the coil; a controller configured to generate a threshold current based on information about a coil current flowing in the coil and an input voltage applied to the coil, and to generate a clock signal by comparing the coil current with the threshold current; and a switch driver configured to generate a switch driving signal to turn on or off the switching device of the inverter unit by dividing a frequency of the clock signal. 
     Other aspects, advantages, and salient features of the disclosure will become apparent to those skilled in the art from the following detailed description, which, taken in conjunction with the annexed drawings, discloses exemplary embodiments of the disclosure. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other features and advantages of the present disclosure will become more apparent by describing in detail exemplary embodiments thereof with reference to the attached drawings in which: 
         FIG. 1  is a perspective view of an induction heating cooker, according to an embodiment of the present disclosure; 
         FIG. 2  illustrates a structure of a coil of the induction heating cooker of  FIG. 1 ; 
         FIG. 3  illustrates a heating principle of the induction heating cooker of  FIG. 1 ; 
         FIG. 4  a block diagram of a high frequency power supply of the induction heating cooker of  FIG. 1 , according to an embodiment of the present disclosure; 
         FIG. 5  is a circuit diagram of the high frequency power supply of  FIG. 4 ; 
         FIG. 6  is an equivalent circuit of the induction heating cooker of  FIG. 1 ; 
         FIG. 7  illustrates a switching current curve at a high frequency power supply circuit of the induction heating cooler of  FIG. 1 ; 
         FIG. 8  illustrates operations of a resonance point tracking controller as shown in  FIG. 5 ; 
         FIG. 9  is a block diagram of a high frequency power supply of the induction heating cooler of  FIG. 1 , according to another embodiment of the present disclosure; 
         FIG. 10  is a circuit diagram of the high frequency power supply of  FIG. 9 ; 
         FIG. 11  illustrates operations of a resonance point tracking controller as shown in  FIG. 10 ; and 
         FIG. 12  illustrates operations of a current controller as shown in  FIG. 9 . 
     
    
    
     Throughout the drawings, like reference numerals will be understood to refer to like parts, components, and structures. 
     DETAILED DESCRIPTION 
     The present disclosure will now be described more fully with reference to the accompanying drawings, in which exemplary embodiments of the disclosure are shown. The disclosure may, however, be embodied in many different forms and should not be construed as being limited to the embodiments set forth herein; rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the concept of the disclosure to those skilled in the art. Like reference numerals in the drawings denote like elements, and thus their description will be omitted. 
     It will be understood that, although the terms first, second, third, etc., may be used herein to describe various elements, components, regions, layers and/or sections, these elements, components, regions, layers and/or sections should not be limited by these terms. These terms are only used to distinguish one element, component, region, layer or section from another region, layer or section. Thus, a first element, component, region, layer or section discussed below could be termed a second element, component, region, layer or section without departing from the teachings of the present disclosure. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. It is to be understood that the singular forms “a,” “an,” and “the” include plural references unless the context clearly dictates otherwise. 
     The term “include (or including)” or “comprise (or comprising)” is inclusive or open-ended and does not exclude additional, unrecited elements or method steps. “Unit”, “module”, “block”, etc. used herein each represent a unit for handling at least one function or operation, and may be implemented in hardware, software, or a combination thereof. 
     In the following description, various embodiments of an induction heating apparatus are provided with reference to  FIGS. 1 to 12 . While embodiments of  FIGS. 1 to 12  illustrate an induction heating cooker as an example of an induction heating apparatus, the embodiments of the present disclosure are not limited to the cooker, but may also be applied in various industrial fields that use induction heating, such as a heating apparatus for heating a toner in a fuser of a printer, a heat processing apparatus for metal surface reinforcement. 
       FIG. 1  is a perspective view of an induction heating cooker, according to an embodiment of the present disclosure. Referring to  FIG. 1 , the induction heating cooker  1  may include a main body  1 . On the top of the main body  1 , a cooking plate  2  may be prepared to have a cooking vessel  52  put thereon. The cooking vessel  2  may be made of ceramic materials. Inside the main body  1  and under the cooking plate  2 , a plurality of coils  54  may be prepared to serve heat sources for the cooking plate  2 . The coils  54  may be equally spaced under and across the entire area of the cooking plate  2 . In the embodiment of  FIG. 1, 16  coils  54  are arranged in 4×4 matrix form. However, unlike this embodiment, the coils  54  may not be equally spaced under and across the entire area of the cooling plate  2  in other embodiments. Under the cooking plate, a control unit  3  is also prepared to drive the coils  54 . A control panel  4  is prepared atop the main body  1  and may include an input unit  80  for receiving a command from a user and delivering the command for the control unit  3  to drive the coils  54  and a display  90  for displaying information regarding works of the induction heating cooker. 
       FIG. 2  illustrates a structure of a coil of the induction heating cooker of  FIG. 1 . Referring to  FIG. 2 , one of the coils (briefly called ‘the coil  54 ’) is prepared under the cooking plate  2  in a helical form and electrically connected to a high frequency power supply  206 . The high frequency power supply  206  may apply high frequency current to the coil  54 . 
     A cooking vessel  52  containing some food may be put on the top of the cooking plate  2 . When the high frequency current is applied to the coil  54  while the induction heating cooker is in operation, magnetic fields are produced in a direction as represented by  202  and the magnetic fields cause induced current, i.e., eddy current as represented by  204  to be generated on the bottom of the cooking vessel  52  on the cooking plate  2 . 
       FIG. 3  illustrates a heating principle of the induction heating cooker of  FIG. 1 . The induction heating cooker heats the food inside the cooking vessel  52  using electric resistance and the eddy current generated by the Faraday&#39;s law. 
     Turning back to  FIG. 2 , it has been discussed that the high frequency current flowing in the coil  54  produces magnetic fields around the coil  54  and the magnetic fields induces the eddy current on the bottom of the cooking vessel  52 . The high frequency current may have a frequency in a range from about 20 kHz to about 35 kHz. If the cooking vessel  52  made of a metal is located within a coverage affected by the magnetic fields produced around the coil  54 , the magnetic fields penetrates the bottom of the cooking vessel  52  and induces the eddy current by the Faraday&#39;s law. Interaction of the eddy current and the electric resistance of the cooking vessel  52  leads to Joule heating of the cooking vessel  52 , which may heat the food contained in the cooking vessel  52 . In this regard, since the cooking vessel  52  serves as a heating source itself, it may be made of a metal material, such as iron, stainless steel, nickel, etc. However, aluminum, ceramic, or glass may not suitable for the cooking vessel  52  used with the induction heating apparatus because such materials have little electrical resistance and are not easily heated. 
       FIG. 4  a block diagram of a high frequency power supply of the induction heating cooker of  FIG. 1 , according to an embodiment of the present disclosure. Referring to  FIG. 4 , an inverter unit  402  may include the coil  54  as shown in  FIGS. 2 and 3 . When Alternate Current (AC) power from an AC power source  404  is supplied to the coil  54  of the inverter unit  402  through a filter unit  406  and a rectifier  408 , a resonance point tracking controller  410  (or called a ‘first controller’) and a switch driver  412  may be involved in supplying power to the inverter unit  402  in order to minimize a power loss that is likely to occur in the inverter unit  402 . 
     As shown in  FIG. 4 , the AC power source  404  may supply common AC power with e.g., a voltage of 325V and a frequency of 50 Hz. The common AC power supplied from the AC power source  404  may go through noise cancellation in the filter unit  406 , and then delivered to the rectifier  408 . The rectifier  408  may rectify the AC power to DC power, and the DC power is delivered to the inverter unit  402 . The inverter unit  402  may convert the DC power to high frequency (AC) power by switching operations, and the high frequency power is applied to the coil  54 . 
     The resonance point tracking controller  410  may control the switch driver  412  to minimize the power loss by optimizing the switching operations for supplying power to the coil  54  even if some electrical parameters (e.g., equivalent inductance and equivalent resistance) of the coil  54  change. The switch driver  412  may generate and provide switch driving signals P 1  and P 2  for the inverter unit  402  to perform switching operations. The inverter unit  402  may perform the switching operations with the switch driving signals P 1  and P 2 . 
       FIG. 5  is a circuit diagram of the high frequency power supply of  FIG. 4 . Referring to  FIG. 5 , the filter unit  406  may include transformers and capacitors and may cancel noise mixed into the power being supplied from the AC power source  404 . The rectifier  408  may constitute a bridge rectifier circuit of multiple diodes. The AC power output from the filter unit  406  may be rectified into DC power by rectification of the multiple diodes of the rectifier  408 . 
     The inverter unit  402  may be based on a half-bridge circuit including multiple switching devices Q 1  and Q 2  and multiple capacitors C r1  and C r2 . The switching devices Q 1  and Q 2  may be turned on or turned off by the switch driving signals P 1  and P 2  generated by the switch driver  412 . The switching devices Q 1  and Q 2  may be Insulated Gate Bipolar Transistors (IGBTs). In the inverter unit  402 , input voltage V i  may be equally divided by the capacitors C r1  and C r2  with approximately the same capacitance and the divided voltage V i /2 appears across each of the capacitors C r1  and C r2 . Feedback diodes may be connected across the switching devices Q 1  and Q 2  to have current continuously flow in an inductive load. Alternately turning on and off the switching devices Q 1  and Q 2  in the inverter unit  402  may enable certain frequency AC current to flow in the coil  54 . A current sensor  502  may be located in a current path between the coil  54  and the capacitors C r1  and C r2 . The current sensor  502  may detect a magnitude of the current flowing in the coil  54 , i.e., coil current I L . Information about the coil current I L  (e.g., information about the magnitude of the coil current I L ) may be provided to the resonance point tracking controller  410 . Furthermore, information about the voltage applied to the coil  54  (e.g., information about the input voltage V, across the two capacitors C r1  and C r2  in series) may be applied to the resonance point tracking controller  410 . 
     In the inverter unit  402 , a frequency of the high frequency current applied to the coil  54  may be fixed or have a particular value according to user settings. Since the frequency of the high frequency current determines strength of magnetic fields around the coil  54  and an inductive current is created in proportion to the strength of the magnetic fields, a heating value in the cooking vessel  52  may be finally determined in proportion to the frequency of the high frequency current applied to the coil  54 . 
     The resonance point tracking controller  410  may control a comparator  504  (or called a first comparator) to compare the information about the coil current I L  with the information about the input voltage V i , and generate a clock signal CLK with a frequency determined by the comparison. An absolute value of the coil current I L  input to the resonance point tracking controller  410  may be input to an inverting input (−) of the comparator  504 . A threshold current I th  (or called a first threshold current) may be input to a non-inverting input (+) of the comparator  504 . Accordingly, if the absolute value of the coil current I L  is greater than the threshold current I th , a low level of the clock signal CLK may be output from the comparator  504 , forming a low level interval; and otherwise if the absolute value of the coil current I L  is less than or equal to the threshold current I th , a high level of the clock signal CLK may be output from the comparator  504 , forming a high level interval. The threshold current I th  input to the non-inverting input (+) of the comparator  504  may be determined by a peak value I LPK  of the coil current I L  divided by the input voltage V i , i.e., I LPK /V i . A value resulting from multiplication of the threshold current I th  by a proportional factor K may be input to the non-inverting input (+) of the comparator  504 . Detailed operations of the resonance point tracking controller  410  may be discussed later in connection with  FIG. 8 . 
     The switch driver  412  may generate multiple switch driving signals P 1  and P 2  by dividing the clock signal CLK generated by the resonance point tracking controller  410 . The multiple switch driving signals P 1  and P 2  may turn on or turn off the switching devices Q 1  and Q 2  of the inverter unit  402  as discussed above. The clock signal CLK generated by the resonance point tracking controller  410  may be input to a D flip-flop  506  of the switch driver  412  as a clock signal. A logic Input D of the D flip-flop  506  is connected to a negative logic output Q′ of the D flip-flop  506 . The frequency of the clock signal CLK generated by the resonance point tracking controller  410  may be divided by 2. A positive logic signal output from a positive logic output Q of the D flip-flop  506  may go through a delay circuit  508  and an AND gate  510  and be output as the switch driving signal P 1  to turn on or turn off the switching device Q 1 , and a negative logic signal output from a negative logic output Q′ of the D flip-flop  506  may go through a delay circuit  518  and an AND gate  520  and be output as the switch driving signal P 2  to turn on or turn off the switching device Q 2 . Due to the work of the D flip-flop  506 , the two switch driving signals P 1  and P 2  have opposite phases. Due to the work of the delay circuits  508  and  518 , dead time may occur between the two switch driving signals P 1  and P 2 , which prevents overlapping of the two switch driving signals P 1  and P 2 , thereby avoiding a situation where the two switching devices Q 1  and Q 2  of the inverter unit  402  are turned on at the same time. 
       FIG. 6  is an equivalent circuit of the induction heating cooker of  FIG. 1 . The induction heating cooker may be represented with a transformer equivalent model having the coil  54  and a load, e.g., the cooking vessel  52  for a primary side and a secondary side, respectively. As shown in  FIG. 6 , the equivalent model may be represented with equivalent inductance Leq and equivalent resistance Req in series. The equivalent circuit may be expressed in equation 1 as follows:
 
 V   1 =( R   1   +jwL   1 ) I   1   −jwMI   2  
 
0=− jwMI   1 +( R   2   +jwL   2 ) I   2   (1)
 
     In equation 1, M refers to mutual inductance. With respect to I 2 , equation 1 may be re-arranged into equation 2 as follows: 
     
       
         
           
             
               
                 
                   
                     I 
                     2 
                   
                   = 
                   
                     
                       j 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         wMI 
                         1 
                       
                     
                     
                       
                         R 
                         2 
                       
                       + 
                       
                         j 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         
                           wL 
                           2 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
     By equations 1 and 2, parameters of the equivalent circuit may be expressed in equation 3 as follows: 
     
       
         
           
             
               
                 
                   
                     
                       V 
                       1 
                     
                     
                       I 
                       1 
                     
                   
                   = 
                   
                     
                       R 
                       eq 
                     
                     + 
                     
                       j 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         wL 
                         eq 
                       
                     
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
     The equivalent parameters Leq and Req may vary with size and location of a load for heating (e.g., the cooking vessel  52 ), a distance between the coil  54  and the load, conductivity and permeability of the load, operating frequency, etc. To control current of the coil  54  that is robust to variation of the parameters Leq and Req, a switching frequency Fsw of the switching device Q 2  may be higher than a resonance frequency Fr. To control current of the coil  54  that is robust to variation of the parameters Leq and Req, a phase difference φ between the voltage and the current applied to the coil  54  may be great. However, the greater the phase difference φ becomes, the greater the turn-off current I off  of the switching devices Q 1  and Q 2  becomes, thereby increasing switching losses. 
       FIG. 7  illustrates a switching current curve at a high frequency power supply circuit of the induction heating cooler of  FIG. 1 . Referring to  FIG. 7 , when the switching devices Q 1  and Q 2  are repeatedly turned on and turned off, the greater the phase difference φ between a drain-source voltage V DS  and a collector-emitter current (I CE , which is the same as the coil current I L ) of the switching devices Q 1  and Q 2 , the greater the switching loss. Accordingly, the embodiment of the present disclosure aims at minimizing the phase difference φ between the voltage and the current to minimize the switching loss. 
       FIG. 8  illustrates operations of the resonance point tracking controller  410  as shown in  FIG. 5 . Turning back to  FIG. 5 , a configuration of the resonance point tracking controller  410  of the induction heating cooker has been described.  FIG. 8  illustrates how the resonance point tracking controller  410  minimizes the phase difference φ. 
     Referring to  FIG. 8 , intervals where the absolute value of the coil current I L  is greater than the threshold current I th  may correspond to low level intervals of the clock signal CLK, while intervals where the absolute value of the coil current I L  is equal to or less than the threshold current I th  correspond to high level intervals of the clock signal CLK. It may be seen that the phase difference φ between the drain-source voltage V DS1  or V DS2  and the coil current I L  of the switching devices Q 1  and Q 2  decreases because the less the threshold current I th , the shorter the high level interval. It is desirable to set the threshold current I th  to be proportional to I LPK /V i  so as to keep the turn-off current for the switching devices Q 1  and Q 2  minimum. Here, I LPK  refers to a peak value of the coil current I L  and V i  refers to an input voltage. 
       FIG. 9  is a block diagram of a high frequency power supply of the induction heating cooler of  FIG. 1 , according to another embodiment of the present disclosure. Referring to  FIG. 9 , an inverter unit  902  may include the coil  54  as shown in  FIGS. 2 and 3 . When AC power from an AC power source  904  is supplied to the coil  54  of the inverter unit  902  through a filter unit  906  and a rectifier  908 , a resonance point tracking controller  910  (or called a ‘first controller’), a current controller  914  (or called a ‘second controller’), and a switch driver  912  may be involved in supplying power to the inverter unit  902  in order to minimize a power loss that is likely to occur in the inverter unit  902 . 
     As shown in  FIG. 9 , the AC power source  904  may supply common AC power with e.g., a voltage of 325V and a frequency of 50 Hz. The common AC power supplied from the AC power source  904  may go through noise cancellation in the filter unit  906 , and then delivered to the rectifier  908 . The rectifier  908  may rectify the AC power to DC power, and the DC power is delivered to the inverter unit  902 . The inverter unit  902  may convert the DC power to high frequency (AC) power by switching operations, and the high frequency power is applied to the coil  54 . 
     The resonance point tracking controller  910  may control the switch driver  912  to minimize the power loss by optimizing the switching operations for supplying power to the coil  54  even if some electrical parameters (e.g., equivalent inductance and equivalent resistance) of the coil  54  change. The switch driver  912  may generate and provide switch driving signals P 1  and P 2  for the inverter unit  902  to perform switching operations. The inverter unit  902  may perform the switching operations with the switch driving signals P 1  and P 2 . 
     The current controller  914  is to control an average current in the coil  54  through low frequency switching operations of Pulse Width Modulation (PWM). Specifically, the current controller  914  may receive information about an absolute value (magnitude) of the coil current I L  from the resonance point tracking controller  910  and then generate an enable (EN) signal that enables the clock signal CLK output from the resonance point tracking controller  910  to be input to the switch driver  912 . 
       FIG. 10  is a circuit diagram of the high frequency power supply of  FIG. 9 . Referring to  FIG. 10 , the filter unit  906  may include transformers and capacitors and may cancel noise mixed into the power being supplied from the AC power source  904 . The rectifier  908  may constitute a bridge rectifier circuit of multiple diodes. The AC power output from the filter unit  906  may be rectified into DC power by rectification of the multiple diodes of the rectifier  908 . 
     The inverter unit  902  may be based on a half-bridge circuit including multiple switching devices Q 1  and Q 2  and multiple capacitors C r1  and C r2 . The switching devices Q 1  and Q 2  may be turned on or turned off by the switch driving signals P 1  and P 2  generated by the switch driver  912 . The switching devices Q 1  and Q 2  may be Insulated Gate Bipolar Transistors (IGBTs). In the inverter unit  902 , input voltage V; may be equally divided by the capacitors C r1  and C r2  with approximately the same capacitance and the divided voltage V i /2 appears across each of the capacitors C r1  and C r2 . Feedback diodes may be connected across the switching devices Q 1  and Q 2  to have current continuously flow in an inductive load. Alternately turning on and off the switching devices Q 1  and Q 2  in the inverter unit  902  may enable certain frequency AC current to flow in the coil  54 . A current sensor  1002  may be located in a current path between the coil  54  and the capacitors C r1  and C r2 . The current sensor  1002  may detect a magnitude of the current flowing in the coil  54 , i.e., coil current I L . Information about the coil current I L  (e.g., information about the magnitude of the coil current I L ) may flow to the resonance point tracking controller  910 . Furthermore, information about the voltage applied across the coil  54  (e.g., information about the input voltage V i  across two capacitors C r1  and C r2  in series) may be applied to the resonance point tracking controller  910 . 
     In the inverter unit  902 , a frequency of the high frequency current applied to the coil  54  may be fixed or have a particular value according to user settings. Since the frequency of the high frequency current determines strength of magnetic fields around the coil  54  and an inductive current is created in proportion to the strength of the magnetic fields, a heating value in the cooking vessel  52  may be finally determined in proportion to the frequency of the high frequency current applied to the coil  54 . 
     The resonance point tracking controller  910  may control a comparator  1004  (or called a first comparator) to compare the information about the coil current I L  with the information about the input voltage V i , and generate a clock signal CLK with a frequency determined by the comparison. An absolute value of the coil current I L  input to the resonance point tracking controller  910  may be input to an inverting input (−) of the comparator  1004 . A threshold current I th  (or called a first threshold current) may be input to a non-inverting input (+) of the comparator  1004 . Accordingly, if the absolute value of the coil current I L  is greater than the threshold current I th , a low level of the clock signal CLK may be output from the comparator  1004 , forming a low level interval; and otherwise if the absolute value of the coil current I L  is less than or equal to the threshold current I th , a high level of the clock signal CLK may be output from the comparator  504 , forming a high level interval. The threshold current I th  input to the non-inverting input (+) of the comparator  1004  may be determined by a peak value I LPK  of the coil current I L  divided by the input voltage V i , i.e., I LPK /V i . A value resulting from multiplication of the threshold current I th  by a proportional factor K may be input to the non-inverting input (+) of the comparator  1004 . Detailed operations of the resonance point tracking controller  910  may be discussed later in connection with  FIG. 11 . 
     The current controller  914  may receive information about an absolute value (magnitude) of the coil current I L  from the resonance point tracking controller  910  and a reference current I LREF  (or called a second threshold current), perform Proportional Integral (PI) control over a difference between the absolute value of the coil current I L  and the reference current I LREF , and ensure that a PI control value P ref  resulting from the PI control is input to the non-inverting input (+) of a comparator  1058  (or called a second comparator). Low frequency (e.g., 1 Hz) sawtooth waves  1056  may be input to the inverting input (−) of the comparator  1058 . Here, the reference current I LREF  refers to a current value that represents a target output of the induction heating cooker set by the user. The current controller  914  may generate a pulse signal as an enable (EN) signal having a high level value in an interval where the PI control value P ref  is greater than the sawtooth wave and a low level value in an interval where the PI control value Pref is equal to or less than the sawtooth wave. The enable signal EN may be input to an AND gate  1060  of the switch driver  912 . 
     The switch driver  912  may generate multiple switch driving signals P 1  and P 2  by dividing the clock signal CLK generated by the resonance point tracking controller  910 . The multiple switch driving signals P 1  and P 2  may turn on or turn off the switching devices Q 1  and Q 2  of the inverter unit  902  as discussed above. The clock signal CLK generated by the resonance point tracking controller  910  may also be input to the AND gate  1060  of the switch driver  912 . Thus, the clock signal CLK and the enable signal EN may be input signals for the AND gate  1060 . For the high level interval of the enable signal EN, the intact clock signal CLK is output from the AND gate  1060 ; and for the low level interval of the enable signal EN, only a low level signal may be output from the AND gate  1060 . After all, it may be seen that the enable signal EN enables the clock signal CLK output from the resonance point tracking controller  910  to be selectively sent to a D-flip-flop  1006  of the switch driver  912 . The output signal from the AND gate  1060  may be a clock signal for the D flip-flop  1006 . A logic Input D of the D flip-flop  1006  is connected to a negative logic output Q′ of the D flip-flop  1006 . The frequency of the clock signal CLK generated by the resonance point tracking controller  910  (more specifically, the frequency of a signal output from the AND gate  1060 ) may be divided by 2. A positive logic signal output from a positive logic output Q of the D flip-flop  1006  may go through a delay circuit  1008  and an AND gate  1010  and be output as the switch driving signal P 1  to turn on or turn off the switching device Q 1 , and a negative logic signal output from a negative logic output Q′ of the D flip-flop  1006  may go through a delay circuit  1018  and an AND gate  1020  and be output as the switch driving signal P 2  to turn on or turn off the switching device Q 2 . Due to the work of the D flip-flop  1006 , the two switch driving signals P 1  and P 2  have opposite phases. Due to the operation of the delay circuits  1008  and  1018 , dead time may occur between the two switch driving signals P 1  and P 2 , which prevents overlapping of the two switch driving signals P 1  and P 2 , thereby avoiding a situation where the two switching devices Q 1  and Q 2  of the inverter unit  902  are turned on at the same time. 
       FIG. 11  illustrates operations of the resonance point tracking controller  910  as shown in  FIG. 10 . Turning back to  FIG. 10 , configuration of the resonance point tracking controller  910  of the induction heating cooker has been described.  FIG. 11  illustrates how the resonance point tracking controller  910  minimizes the phase difference φ. 
     Referring to  FIG. 11 , intervals where the absolute value of the coil current I L  is greater than the threshold current I th  may correspond to low level intervals of the clock signal CLK, while intervals where the absolute value of the coil current I L  is equal to or less than the threshold current I th  correspond to high level intervals of the clock signal CLK. It may be seen that the phase difference φ between the drain-source voltage V DS1  or V DS2  and the coil current I L  of the switching devices Q 1  and Q 2  decreases because the less the threshold current I th , the shorter the high level interval. It is desirable to set the threshold current I th  to be proportional to I LPK /V i  so as to keep the turn-off current for the switching devices Q 1  and Q 2  minimum. Here, I LPK  refers to a peak value of the coil current I L  and V i  refers to an input voltage. 
       FIG. 12  illustrates operations of the current controller  914  as shown in  FIG. 9 . Referring to  FIG. 12 , the current controller  914  may compare a PI control value P ref  resulting from PI control with a low frequency (e.g., about 1 Hz) sawtooth wave  1056 , and generate a pulse signal as an enable (EN) signal having a high level value in an interval where the PI control value Pref is greater than the sawtooth wave and a low level value in an interval where the PI control value P ref  is equal to or less than the sawtooth wave. The switch driver  912  may ensure that the voltage V i  and the coil current are applied to the coil  54  only in high level intervals of the enable signal EN, as shown in the bottom of  FIG. 12 , by turning on and off the switching devices Q 1  and Q 2  with the switch driving signals P 1  and P 2  generated by dividing the clock signal CLK only in the high level interval of the enable signal EN. After all, it may be seen that quantities of the voltage V i  and the coil current I L  may depend on the size of the high level interval (i.e., the pulse width) of the enable signal EN. 
     If the magnitude of the reference current I LREF  is large and thus leads to increase in the PI control value P ref , the width of the high level interval of the enable signal becomes larger and thus greater voltage V i  and coil current I L  may be applied to the coil  54 ; but otherwise, if the magnitude of the reference current I LREF  is small and thus leads to decrease in the PI control value P ref , the width of the high level interval of the enable signal becomes smaller and thus relatively less magnitude of voltage V i  and coil current I L  may be applied to the coil  54 . As such, it may be seen that quantities of the voltage V i  and the coil current I L  to be applied to the coil  54  are controlled based on the quantity of the reference current I LREF  that represents the target output set by the user while the frequency of the voltage V i  and the coil current I L  remains the same. While the plurality of coils  54  of the induction heating cooker are operating, if respective frequencies of the coils  54  are different, the difference in those frequencies may cause noise. The embodiments of the present disclosure may have an effect of noise reduction by controlling only conducting intervals by pulse width control as shown in  FIG. 12  while keeping the frequency of the voltage V i  and the coil current I L  for each of the plurality of coils  54  intact. 
     Several embodiments have been described in connection with e.g., mobile communication terminals, but a person of ordinary skill in the art will understand and appreciate that various modifications can be made without departing the scope of the present disclosure. Thus, it will be apparent to those ordinary skilled in the art that the disclosure is not limited to the embodiments described, which have been provided only for illustrative purposes. 
     Although a few embodiments have been shown and described, it would be appreciated by those skilled in the art that changes may be made in these embodiments without departing from the principles and spirit of the invention, the scope of which is defined in the claims and their equivalents.