Patent Publication Number: US-8971718-B2

Title: Down-sampling clock and data recovery circuit having selectable rate and phase output and method of operation thereof

Description:
TECHNICAL FIELD 
     This application is directed, in general, to clock and data recovery (CDR) circuits and, more specifically, to a down-sampling CDR circuit having a selectable rate and phase output and a method of down-sampling with a selectable rate and phase output. 
     BACKGROUND 
     A CDR circuit is a key building block in many high speed wireline data communication applications, such as optical networks, backplane interconnects and chip-to-chip interconnection. Its function is to extract a transmitted data sequence and clocking/timing information from a received signal, which has typically been subjected to noise and signal distortion. The CDR circuit detects signal transitions in the received data and produces a stable clock signal (namely, a recovered clock signal). The recovered clock signal drives a decision circuit that samples the received signal and produces a retimed data stream. A deserializer circuit follows the CDR circuit to convert the recovered bit stream into word-aligned data. 
     The CDR circuit, together with the deserializer circuit, plays an important role in the overall performance of a high speed transmission system. Not only does it determine bit error rate (BER) and the stability of the transmission link, but also consumes a substantial portion of overall receiver power. 
     SUMMARY 
     One aspect provides a CDR circuit. In one embodiment, the CDR circuit includes: (1) a line rate CDR circuit having a voltage controlled oscillator, the line rate CDR circuit configured to recover a raw data stream at a receiving line rate, (2) a fixed-rate down-sampler coupled to the line rate CDR circuit and configured to down-sample the raw data stream based on a fixed-rate and (2) a variable-rate down-sampler coupled to the fixed-rate down-sampler and configured further to down-sample the raw data sample based on a variable-rate. 
     Another aspect provides a method of recovering a clock and data from a received raw data stream. In one embodiment, the method includes: (1) recovering a raw data stream at a receiving line rate using a voltage controlled oscillator, (2) initially down-sampling the raw data stream based on a fixed rate and (3) further down-sampling the raw data sample based on a variable rate. 
     Yet another aspect provides a bit-interleaving passive optical network (BI-PON) optical network transceiver (ONT) receiver front-end. In one embodiment, the receiver front-end includes a CDR circuit, including: (1) a line rate CDR circuit having a voltage controlled oscillator, the line rate CDR circuit configured to recover a raw data stream at a receiving line rate, (2) a fixed-rate down-sampler coupled to the line rate CDR circuit and configured to down-sample the raw data stream based on a fixed rate and (3) a variable-rate down-sampler coupled to the fixed-rate down-sampler and configured further to down-sample the raw data sample based on a variable rate. 
    
    
     
       BRIEF DESCRIPTION 
       Reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a block diagram of one embodiment of a CDR circuit with selectable phase and rate output; 
         FIG. 2  is a schematic diagram of one embodiment of an integrated voltage-controlled oscillator (VCO); 
         FIG. 3  is a schematic diagram of one embodiment of an active differential capacitor bank; 
         FIG. 4  is a diagram of a dual-loop control method for use with the CDR circuit embodiment of  FIG. 1 ; 
         FIG. 5  is a block diagram of one embodiment of a fixed-rate down-sampler; 
         FIG. 6  is a block diagram of one embodiment of a variable-rate down-sampler; and 
         FIG. 7  is a schematic diagram of one embodiment of a BI-PON ONT receiver front-end. 
     
    
    
     DETAILED DESCRIPTION 
     As stated above, a CDR circuit&#39;s function is to extract a transmitted data sequence and clocking/timing information from a received signal by detecting signal transitions in the received data and recovering a stable clock signal (or, simply, “clock”) from them. The recovered clock drives a decision circuit that samples the received signal and produces a retimed data stream. 
     In high speed, multi-gigabit receiver design, performance is generally determined by the maximum amount of jitter a receiver can tolerate. Since the received signal is retimed by latching the received (input) signal with the recovered clock, bit errors occur when either the clock or data is sampled too early or too late. To obtain a stable recovered clock, a clock recovery circuit is usually based on a phase-locked loop (PLL), which includes a voltage-controlled oscillator (VCO). 
     A high performance CDR circuit typically decomposes the VCO control into fine and coarse tuning, acquiring frequency and phase in two sequential steps, or loops. Often using a single VCO, such circuit first enables a frequency tracking loop to lock to a reference clock and monitors the frequency error between the loop frequency and the reference clock. When the frequency error drops to a low threshold value, a lock detector associated with the PLL switches to a phase tracking loop. The phase tracking loop fine tunes the VCO such that the CDR circuit output locks to the received data. Even after the CDR circuit has locked to the incoming data, the lock detector continues to operate. The lock detector switches back to the frequency tracking loop if the phase tracking loop loses lock, usually due to unexpected noise. 
     The foregoing circuit works well but consumes substantial power. Unfortunately, a major challenge looms in trying to reduce the power consumed by a high performance CDR circuit, stemming from the use of low supply voltages. As the supply voltage is reduced, the tuning range of a VCO is likewise reduced. Consequently, for a given ripple on its control line, the VCO suffers from a greater jitter. 
     Described herein are various embodiments of a high performance CDR circuit that exhibits substantially less jitter. Certain embodiments of the novel CDR circuit disclosed herein employ an active differential capacitor bank that not only extends, perhaps greatly, the tuning range of an integrated VCO, but is also particularly suitable for an integrated VCO circuit. To reduce the overall power consumption of the CDR circuit, a power-saving loop control method. 
     Furthermore, conventional CDR circuits typically provide only a fixed-rate data and a recovered clock matching the signaling rate at the line input. Certain embodiments of the novel CDR circuit disclosed herein feature a selectable phase and data rate output for the recovered data. A selectable phase and rate output is especially useful for receiver design requiring a flexible subsampling/decimation capability. As will be evident from the following disclosure, the illustrated embodiments of the novel CDR circuit are simpler and consume less power than conventional CDR circuits. 
       FIG. 1  shows a simplified block diagram of one embodiment of a novel CDR circuit  100 . The CDR circuit  100  includes a line rate CDR circuit  110 , a fixed-rate by-2 m  down-sampler  120  and a variable-rate by-2 n  down-sampler  130 . The line rate CDR circuit  110  employs a dual-loop CDR circuit architecture having dual (frequency and phase) tracking loops as described above. The line rate CDR circuit  110  recovers a raw data stream Rx at its receiving line rate R. A reference clock RefClk is chosen such that the receiving line rate R is a multiple of the frequency of the reference clock. As an example, for a Rx line rate of 10 GHz, a reference clock of 78.125 MHz may be chosen with a divider of 128. Further details regarding some embodiments of the line rate CDR circuit  110  will be presented below, along with two features employed with respect to certain embodiments of the line rate CDR circuit  110  that offer the benefit of lower power consumption and better jitter tolerance. 
     Once the raw data is recovered from the line, it is then sampled by the fixed-rate down-sampler  120 , reducing the raw data rate from the line data rate R down to a lower data rate R/2 m , matching the general range of the final user selectable rates. The fixed-rate down-sampler  120  includes a phase select input PhSel(m) that determines which phase of the resulting data output should be presented. Further details regarding some embodiments of the fixed-rate down-sampler  120  will be presented below. 
     After the fixed-rate down-sampling, the recovered data is sampled again using the variable-rate down-sampler  130 . The variable-rate down-sampler  130  is configured such that a target down-sampling rate and its corresponding phase offset of the recovered data may be selected within a designated range using respective rate set and phase set inputs RateSel and PhSel(n). Further details regarding some embodiments of the variable-rate down-sampler  130  will be presented below. 
     An integrated, LC-based VCO has recently been widely adopted for many applications because of its low noise characteristics and relatively simple and compact structure. (In the context of this disclosure, “integrated” means able to be fabricated into an integrated circuit.)  FIG. 2  illustrates an integrated VCO  200 . The VCO includes a resonance element  210  and an amplifier stage  220 . The resonance element  210 , typically a circuit including inductive and capacitive elements (an “LC tank”), sets the oscillation frequency. Due to the losses in the resonance element  210 , the amplifier stage  220  is employed to provide enough gain to sustain oscillation. The frequency of the oscillation is controlled by a varactor, which implements the capacitance the LC tank, with a control voltage Vc. In a fully integrated VCO, the tuning range of the oscillation frequency should be large enough to cover anticipated fabrication process variations and temperature fluctuations. However, the tuning range provided by a simple varactor is typically limited. Unfortunately, increasing sensitivity of the frequency to the control voltage tends to increase phase noise. Therefore, to extend the tuning range, a fixed capacitor bank  230  can be added to adjust the resonant frequency coarsely, while fine tuning is then achieved by means of varactors Cvar 1  and Cvar 2 . 
     An idealized differential capacitor bank employs a switch coupled between a pair of capacitors, as  FIG. 2  shows (e.g., an unreferenced switch between capacitors C 1 , C 2 ). practical differential capacitor bank uses a metal-oxide-semiconductor field-effect transistor (MOSFET) to implement the switch. However, implementing the switch with a MOSFET introduces two non-ideal characteristics that greatly affect the performance of the VCO. First, a MOSFET has a non-zero ON resistance. This additional resistance affects the quality factor (Q-factor) of the LC tank. To keep the phase noise low, a large MOSFET operating in its triode region is usually required to hold the ON resistance to an acceptable level. Unfortunately, a large MOSFET not only makes the VCO bulky, but even worse has parasitic capacitance that contributes to overall capacitance when the MOSFET is in the OFF state. Such non-negligible capacitance at the OFF state limits the tuning range of the VCO. To reduce this parasitic capacitance, the MOSFET size needs to be reduced substantially. 
     A differential capacitor bank may be improved by using two MOSFET switches instead of one. In this way, the overall ON resistance can be reduced by half given the same MOSFET size. The MOSFET size can thus be decreased to reduce the impact of the parasitic capacitance while maintaining the Q-factor of the LC tank. Two bias resistors are then typically used to provide DC voltage. The DC voltage is set to reduce the MOSFET ON resistance by increasing the overdrive voltage. Unfortunately, the value of the bias resistor has to be made very large to avoid deterioration of the Q-factor of the LC tank. As those skilled in the pertinent art are aware, a compact, high-value resistor is difficult to implement in an integrated circuit. 
       FIG. 3  illustrates one embodiment of a schematic diagram of an active differential capacitor bank  300  that addresses these issues. The active differential capacitor bank  300  uses inverters I 0  and I 1  to generate bias and switch control voltages respectively. As a replacement for the bias resistors described above, two transmission gates M 1 , M 3  and M 2 , M 4  are used. When a switch M 0  is ON, the MOSFETs in the transmission gates M 1 , M 3  and M 2 , M 4  (working in their triode regions) present a large resistance without introducing significant parasitic capacitance. When the switch M 0  is OFF, the bias voltage is high, bringing the source and drain of the switch M 0  to Vdd, increasing depletion layers around the switch M 0 , and further reducing its parasitic capacitance. Furthermore, using a transmission gate M 1 , M 3  instead of a single MOSFET improves the linearity of the parasitic capacitance, decreasing phase noise, especially at the low frequency offset. 
       FIG. 4  is a diagram of a dual-loop control method  400  for use with the CDR circuit embodiment of  FIG. 1 . A loop switch/power control block  410  incorporates power control functionality. Shown are a frequency tracking loop  420  and a phase tracking loop  430 . As the frequency and phase tracking loops  420 ,  430  work sequentially, the loop switch/power control block  410  operates in two power modes: P 1  and P 2 . During operation of the frequency tracking loop  420  (clock frequency synthesis), the loop switch/power control block  410  enables power mode P 1 . During power mode P 1 , the frequency tracking loop  420  controls the VCO, while unused blocks of the phase tracking loop  430  are powered down. When the clock frequency synthesis is completed, the loop switch/power control block  410  enables power mode P 2 . In power mode P 2 , the phase tracking loop  430  controls the VCO. When the frequency tracking loop is no longer needed, its unused blocks are powered down to save power. 
       FIG. 5  is a block diagram of one embodiment of a fixed-rate down-sampler  500 . The fixed-rate down-sampler  500  includes a multiphase generator  510 , a multiplexer  520  and two flip-flops  530 ,  540 . The multiphase generator  510  implements a modified 2 m  counter, taking the recovered Rx line clock and producing eight divide-by-2 3  output clocks with different phases. Given a PhaseSel input (most significant m bits) the multiplexer  520  selects one of the eight clocks to sample the recovered data stream. The sampled data is then retimed for the next stage. For applications having an extremely high clock frequency at the line rate, e.g., 40 GHz+, the fixed-rate down-sampler  500  may be realized using a higher speed logic circuit with slightly higher power consumption, while other parts of the CDR circuit can be implemented using lower speed logic gates. By doing so, the overall power consumption of the CDR circuit can be kept low. 
       FIG. 6  is a block diagram of one embodiment of a variable-rate down-sampler  600 . The core of the variable-rate down-sampler  600  is a modified n-bit preload binary counter  610 . The n-bit preload binary counter  610  takes the output clock of the previous stage as a clock input. The n-bit preload binary counter  610  begins counting from a preload value which is set by the second (least significant n bits) part of the PhaseSel input. The n-bit preload binary counter  610  produces all (n+1), divide-by-2 to divide-by-2 (n+1 ) rate clock outputs, together with the original input clock. By using a (n+1):1 multiplexer  620 , a target rate output can be selected to sample the recovered data from the previous stage using a flip-flop  630 . 
       FIG. 7  is a schematic diagram of one embodiment of a BI-PON ONT receiver front-end  700 . The front-end  700  implements an embodiment of the CDR circuit disclosed herein having two individually selectable output legs, instead of only one output. The first output leg, “header output,” or “hdr,” has a fixed target rate of divide-by-2 8  but with a fully selectable phase output, i.e., m=3 and n=5. The second output leg, “payload output,” or “pload,” has a selectable sampling rate ranging from divide-by-2 3  to divide-by-2 10  with a fully selectable phase output, i.e., m=3 and n=7. Further details regarding a protocol that can employ the front-end  700 , together with various applications for the front-end  700 , may be found at http://www2.alcatel-lucent.com/blogs/techzine/2012/bipon-a-more-energy-efficient-tdm-pon/. 
     Those skilled in the art to which this application relates will appreciate that other and further additions, deletions, substitutions and modifications may be made to the described embodiments.