Patent Publication Number: US-10766088-B2

Title: Metal working power supply converter system and method

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a Continuation Patent Application of U.S. Non-Provisional application Ser. No. 15/412,620, entitled “Metal Working Power Supply Converter System and Method”, filed Jan. 23, 2017, which is a Continuation Patent Application of U.S. Non-Provisional application Ser. No. 13/925,579, entitled “Metal Working Power Supply Converter System and Method”, filed Jun. 24, 2013, which are both herein incorporated by reference in their entirety for all purposes. 
    
    
     BACKGROUND 
     The invention relates generally to a field of welding systems and systems for performing metal working operations, such as gas metal arc welding (GMAW). More particular, the disclosure relates to innovations in the control of power supplies and converter circuitry used in such systems. 
     Many applications exist for welding and cutting systems used to join two or more workpieces to one another, or to cut workpieces. These applications exist throughout industry, but also for construction, ship building, maintenance, and so forth. In arc welding systems, electrical power is converted to a form useful for a welding process, and the power provides voltage and current necessary to establish and maintain arcs between an electrode and a workpiece. Plasma cutting and similar operations also require conditioned power adapted for the specific process. In gas metal arc welding (GMAW), the arc is established between an electrode that is advanced towards the workpiece and of the workpiece itself. The electrode is consumed insomuch as it is added to the weld as the weld puddle advances during the operation. 
     In welding and cutting power supplies, power electronic circuitry is commonly switched on and off to control the desired power output used for the process. The switching is commonly performed by pulse width modulation (PWM) signals applied to the gates of power electronic switches of converter circuits within the supplies. Conventional systems utilize a single inverter in this converter circuitry, along with an inductor to smooth the output waveform. However, inductors in such systems can be quite large, implying additional cost and weight. Improvements in these systems have included the use of two or more inverters or converters which are switched to provide the desired output. The ripple amplitude of the output current is thereby reduced, consequently reducing the size of the output inductor. 
     One problem in such systems exists insomuch as independently controlled inverter circuits can become unbalanced during operation. The result can be a “walking” deviation in the duty cycles of the two inverters, in which one of the inverters will tend to progressively take on most of the load, causing thermal imbalance, undermining the benefits obtained by the use of multiple inverters. Moreover, current systems may not account for the magnetic dynamics of transformer circuits in the power converters. With common assumptions as to magnetizing and demagnetizing of the transformers, less than the full potential power output is obtained when a PWM duty cycle of 50% is treated as a limit for each of the unipolar or single ended inverter circuits. 
     Improvements are therefore needed in the field that would allow for avoiding drawbacks such as those described above. 
     BRIEF DESCRIPTION 
     The present disclosure sets forth certain solutions that maybe implemented in welding and similar power supplies and that are designed to respond to such needs. In accordance with certain aspects of the present disclosure, a welding or cutting power supply system includes a converter circuit comprising first and second solid state switching circuits each having two power electronic switches for producing output power at a controllable level suitable for a welding or cutting operation, the first and second solid state switching circuits being electrically coupled in to provide combined output. An output inductance is coupled to outputs of both the first and second solid state switching circuits. Control circuitry is coupled to the first and second solid state switching circuits and configured to provide PWM control signals for controlling states of the power electronic switches of the switching circuits to maintain desired loading of the switching circuits independent of the level of the output power. For each cycle of the switching circuits the control circuitry is configured to apply PWM control signals to the first switching circuit based upon a desired duty cycle, to determine the duration of the PWM control signals applied to the first switching circuit, and to apply PWM control signals to the second converter based upon the determined duration without re-determining the duty cycle for the second switching circuit. 
     In accordance with other aspects, a welding or cutting system comprises a power supply comprising first and second solid state switching circuits each having two power electronic switches for producing output power at a controllable level suitable for a welding or cutting operation, the first and second solid state switching circuits being electrically coupled in parallel, and an output inductor coupled to outputs of both the first and second solid state switching circuits. Control circuitry is coupled to the first and second solid state switching circuits and configured to determine PWM control signals for the first switching circuit based upon a desired duty cycle, to determine the duration of the PWM control signals of the first switching circuit, and to determine PWM control signals to the second converter based upon the determined duration without re-determining the duty cycle for the second switching circuit. 
     In accordance with still further aspects, a welding or cutting method, comprises applying PWM control signals to a first switching circuit of a power converter comprising at least two switching circuits operated in an interleaved switching pattern, the control signals of the first switching circuit being based upon a desired duty cycle to produce a welding or cutting output, and determining a duration of the PWM control signals applied to the first switching circuit. PWM control signals are then applied to a second switching circuit of the power converter for a duration based upon the determined PWM duration. 
    
    
     
       DRAWINGS 
       These and other features, aspects, and advantages of the present invention will become better understood when the following detailed description is read with reference to the accompanying drawings in which like characters represent like parts throughout the drawings, wherein: 
         FIG. 1  is a diagrammatical representation of an exemplary GMAW welding system incorporating power conversion circuitry in accordance with the present disclosure; 
         FIG. 2  is a diagrammatical representation of exemplary components of the power conversion circuitry; 
         FIG. 3  is an electrical schematic of a currently contemplated dual two-switch power converter utilizing aspects of the present disclosure; 
         FIG. 4  is a flow chart illustrating exemplary logic for providing control signals to the two converters of the circuit of  FIG. 3 ; 
         FIGS. 5-10  are graphical representations of exemplary control and output waveforms exemplifying the control of the two converters in accordance with the present techniques; 
         FIGS. 11-14  are graphical representations of switched control signals for more fully utilizing the capabilities of the circuitry; and 
         FIGS. 15-19  are graphical representations of exemplary voltage and current waveforms of the type obtainable through use of the control scheme outlined in  FIGS. 11-14 . 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  illustrates an exemplary welding system  10  in the form of a GMAW system, and in particular a MIG welding system. In this exemplary system a power supply  12  receives and converts power that is applied to a wire feeder  14 . The wire feeder delivers the power to a welding torch  16  for completing a welding operation. Although the present disclosure used this GMAW power supply as an exemplary basis for presentation of the new circuitry and control techniques, it should be understood that the same teachings may be applied to power supplies used for other welding processes, as well as for other metal working processes, such as plasma cutting. 
     The power supply  12  receives input power  18  from any suitable source, such as the power grid, an engine generator set, hybrid power supplies, fuel cells, batteries, or a combination of these. Power conversion circuitry  20  converts the power to a form suitable for a welding (or other metal working) process. The power supply may be designed to carry out multiple different welding processes that can be selected by an operator, and the power conversion circuitry includes components, such as solid state switches discussed below, that allow for power conversion in accordance with the desired process. Control and processing circuitry  22  is coupled to the power conversion circuitry  20  and controls the operation of the power conversion circuitry during the selected process. For example, the control and processing circuitry  22  may provide signals that regulate the conductive states of solid state switches within the power conversion circuitry to produce the desired output power, as also discussed below. In many applications the control and processing circuitry will include one or more digital processors or microprocessors with associated memory to store and carry out the processes available on the power supply. Such processes may include constant voltage (CV) processes, constant current (CC) processes, pulsed processes, cutting processes, and so forth. The processes and other welding parameters may be selected via an operator interface  24  that is coupled to the control and processing circuitry  22 . The power supply may further include circuitry that allows for communications with remote or networked components and systems, illustrated as data/network interface  26  in  FIG. 1 . Such circuitry may allow for monitoring of welding operations, logging of weld data, downloading or configuration of new processes and updates to processes, and so forth. Finally, the power supply will sometimes include removable memory  28  that may be used for storing processes, process parameters, system updates, and any suitable data. 
     Power and data may be transferred from the power supply  12  to the wire feeder  14  via one or more cables or cable bundles  30 . The wire feeder itself comprises a drive control circuitry  32  that regulates the operation of a drive assembly  34 . The drive assembly  34  contacts and feeds a wire electrode  36  to the welding operation. The wire electrode is typically stored on a spool  38  within the wire feeder. The wire feeder may also include one or more gas valves for providing shielding gas for a welding operation. Finally, an operator interface  42  may allow certain parameters of the wire feeder to be selected, such as wire feed speed. The power supply and wire feeder may operate in coordination so that wire and gas resources are fed to the welding operation when power is provided for welding at the initiative of the welding operator (e.g., via a control on the torch). In some embodiments the power supply and wire feeder may be integrated into a single package. The wire and gas resources are provided via a weld cable  44  coupled to the torch. A second or work cable  46  is typically clamped or coupled in some manner to a workpiece  48  for completing the electrical circuit. The full circuit is completed during the welding operation by an arc as indicated at reference numeral  50 . 
       FIG. 2  illustrates exemplary functional components for the power conversion circuitry  20 . As shown in  FIG. 2 , this circuitry may include an electromagnetic interference filter  52 . Such circuitry will typically provide an inductive-capacitive filter, and may include protective devices such as varistors. An input rectifier  54  converts AC power to DC power, and will typically include a full bridge rectifier. The DC power is placed on a DC bus, and the circuitry may include bus capacitors to smooth the ripple of the DC waveform on the DC bus. A power factor correction boost circuit  56  may allow for power factor correction with the input power. Some systems may not include this boost circuit. Where provided, this may be a switched circuit that steps up the DC bus voltage to a desired level, such as 440V. An isolated converter  58 , which is discussed in greater detail below, includes, in a presently contemplated embodiment, a dual two-switch interleaved forward converter that itself includes an output rectifier and an output inductor. Exemplary electrical components for this circuitry, along with their operation and control are described below. The output of the converter  58  may be provided to the welding (or other) application. In a presently contemplated embodiment, the PFC boost circuitry  56  and the power converter  58  are provided with control circuits  62  and  64  to regulate their operation. Such control circuits will typically serve to provide gate signals to solid state switches within the power converters in accordance with the particular welding (or cutting) regime, the power output needs, and so forth. 
       FIG. 3  illustrates an exemplary interleaved forward converter of the type mentioned above. The forward converter  66  in this embodiment is a dual two-switch interleaved forward converter that comprises first and second inverter circuits  68  and  70  coupled generally in parallel but controlled in an interleaved fashion. The outputs of the two circuits are joined as illustrated, with an output inductor  72  common to both circuits to smooth the output waveform. It should be noted that in some systems, particularly in welding systems, a welding cable may provide sufficient parasitic inductance such that the separate inductor may not be required, particularly where the frequency of the output waveform is sufficiently high. More particularly, the first inverter circuit  68  comprises two solid state switches  74  and  76 , which may typically be insulated gate bipolar transistors with freewheel diodes  78  and  80 , respectively. Application of pulsed control signals to the gates of these devices allows for creation of an output waveform that is applied to a transformer  82 . Analogous components are provided in the second inverter circuit  70  in the form of switches  84  and  86 , diodes  88  and  90 , and transformer  92 . Each inverter circuit includes diodes  94  and  96  which form an output rectifying circuit, that serve as rectifying and freewheeling functions, respectively. 
     In operation, the two inverters are controlled by pulse width modulated signals that cause the circuits to alternatively produce output that is effectively summed. To ensure that both circuits produce equal output, and that the load is shared, the current control scheme effectively generates and provides pulse width modulated gate drive signals to a first of the inverter circuits, monitors the duration of the “on” state of the first circuit (the period during which the circuit is controlled to produce output power), then pulses the second inverter circuit “on” for a similar period to ensure the same duty cycle. 
     This process is summarized generally in  FIG. 4 . The control logic  98  may be represented as beginning with activation of forward converter one (corresponding to inverter circuit  68  in  FIG. 3 ) as indicated at step  100 . The first forward converter is activated to provide an output for a desired period, depending upon the needs of the welding (or other) operation. In this basic control scenario, each forward converter is allowed to be placed in an “on” output state for one half of the time. Thus, as indicated at step  102 , the “on” period of the first converter is monitored or accumulated, such as by augmenting a counter for clock cycles during which the first converter is “on”. Other techniques may certainly be employed for this purpose, including other methods of measuring time, storing durations of gate drive signals, and so forth. As indicated at step  104 , it is then determined whether the maximum allowed period for activation of the first converter has been reached. In many embodiments, this period may be equal to a 50% duty cycle or less, although periods of greater than 50% may be used, as discussed below. Once the maximum period for activation of the first converter is reached, then, the second forward converter is activated as indicated at step  106 . The second forward converter is activated (e.g., gate drive signals are provided to the solid state switches) for a period equal to that during which the first forward converter was “on”. The query block  108  in  FIG. 4  illustrates that the activation of the second forward converter continues until the period is equal (e.g., the same number of clock cycles counted for the first converter has expired for the second). At step  110  it is determined whether the maximum activation time for the second converter is reached. Once the results of both blocks  108  and  110  are affirmative, the second forward converter is deactivated and the first forward converter may be reactivated 
       FIGS. 5-10  are graphical illustrations of PWM control signals that may be applied to the switches of each of the converters  68  and  70 , as well as output waveforms for currents of each. As illustrated in  FIG. 5 , control signals  118  for the first converter  68  are applied in pulses that may approach a 50% duty cycle. (As described with respect to  FIGS. 11-14 , this duty cycle, may, in fact, exceed 50%.) Thus, “on” periods  120  are those in which the solid state switches  74  and  76  receive gate drive signals to place them in a conductive state (alternatively) and “off” periods  122  are periods during which the signals to the switches maintain them in a non-conductive state. The ratio of the “on” time to the “off” time, designated by reference numerals  124  and  126 , respectively, corresponds to the duty cycle of the first converter  68 . Similarly, as depicted in  FIG. 6 , PWM control signals to the second converter, designated generally by reference numeral  128 , include “on” periods  130  and “off” periods  132  in which the switches  84  and  86  (alternatively) are placed in conductive and non-conductive states, respectively. Here again, as will be appreciated by those skilled in the art, due to the inverter circuitry design the inverter switches are typically switched alternatively. The resulting “on” time  134  and “off” time  136  provide the duty cycle for the second converter  170 . Ideally, these duty cycles will be identical. When combined, then, as illustrated in  FIG. 7 , the interleaved switching of converters  168  and  170  produce a combined duty cycle with “off” periods  140  between the alternating outputs provided by the respective converters. 
     In the present embodiment, as described with respect to  FIG. 4 , each of the converters is available to be placed in a conductive state to provide output power at a specific point in time. Thus, the “on” state  120  in  FIG. 5  for the first converter begins at the left axis in that figure, and the next available point of initiation for placing the first converter in a “on” state begins at the period resulting from the sum of periods  124  and  126  (i.e., the switching period). The second converter similarly is available to be placed in an “on” state at a specific time that is 180° displaced from that of the first converter (i.e., at a point at one half of the switching period). That is, given the total period (period  124 + 126 ), which is equal for both converters, the second converter is available to be switched into the “on” or output state at a time half way through the total switching period. 
       FIG. 8  illustrates the output of the first converter in a current waveform. As illustrated, the current  142  will rise as indicated by reference numeral  144  so long as the converter is being PWM&#39;ed for output, and will then decline following shut-off of the output of the first converter as indicated by reference numeral  146 . The current  148  of the second converter, illustrated in  FIG. 9 , will similarly rise as indicated by reference numeral  150  when this converter is placed in the “on” or output state, and will decline as indicated by reference numeral  152  when “off”. The combined currents, then, illustrated in  FIG. 10  and referred generally by reference numeral  154 , will be the combination of these waveforms exhibiting a reduced ripple current and generally smoothed output at twice the PWM frequency of either converter, as indicated by trace  156 . 
     As described above, the duty cycles of the two converters are maintained equal during this process. To summarize, control of the first converter is based upon the process selected, the output power parameters desired, and so forth. In this way the desired “on” time and waveform characteristics of the first converter are determined, and PWM control signals are applied to the gates of the solid state switches of the first converter for a desired time. This time, designated by reference numeral  124  in  FIG. 5 , is counted, such as by augmenting a counter C (as shown in  FIG. 2 ) within the control circuitry logic to count the number of clock cycles (or any other reference cycle) during which the first converter is commanded to provide output. Subsequently, rather than re-determine the required output for the second converter, the second converter PWM control signals are simply applied for an equal time based upon the counter accumulation which is referenced from the immediately preceding control signals applied to the first converter. The two converters are thus ensured with having the same duty cycle regardless of changes in the duty cycle applied to the first converter. 
     As mentioned above, the present disclosure also provides a mechanism for enhancing the power output of the power converter circuitry described above. In particular, the PWM control signals that command the individual interleaved converters to provide output may be extended beyond the 50% duty cycle range (as discussed below). In conventional systems, it is typically believed that when implementing an inverter topology consisting of a single-ended converter such as a forward converter, the maximum pulse width to the primary of the downstream transformer should not exceed 50%. This traditional treatment is likely the result of the belief that as much time is required to discharge the magnetizing inductance of the transformer as the time required to charge it. Under no-load conditions, this 50% limit generally holds true because the magnetizing inductance charges for the entire time that the primary pulse (“on” period) is applied. However, when the converter is running in constant conduction mode (CCM) and there is load current flowing, there is little or no magnetizing current flowing in the transformer until the current in the leakage inductance matches the reflected load current. Under high load conditions, it may require several microseconds to “charge” the leakage inductance. Also, during this leakage inductance charging time, no secondary voltage is applied to the transformer secondary, which implies that no power is being transformed to the load. This phenomenon may result in an “effective” pulse width where the secondary pulse width is equal to the primary pulse width minus the leakage inductance charge time (which is a function of the load current). When the primary side pulse width is approaching its maximum, say 12.5 microseconds for a 40 KHz forward converter, the secondary pulse width can be substantially less, say on the order of 9 microseconds. This results in an effective pulse width duty of 36% and not 50%. Consequently, if the output voltage is equal to the input voltage times the PWM duty cycle, a limit of the output mean voltage that can be achieved is below the typical 50% limit. 
     In accordance with the present disclosure, a maximum primary pulse width may be allowed to extend beyond the 50% limit based upon the amount of load current. This extension results from a realization that half of the leakage inductance charge time could be added to the pulse width and would still provide adequate time for the magnetizing inductance to discharge without changing the overall cycle period. By way of example, this could raise the effective secondary pulse width to 43% or an additional 7% load voltage without requiring a change in the turns ratio of the transformer. This technique is generally illustrated in  FIGS. 11-14 . 
     As shown in  FIG. 11 , the PWM drive signals  158  for the first converter comprise PWM commands  160  that place the converter in an “on” state, followed by periods  162  in which the converter is “off”. An extension  164  of the “on” period beyond the normal 50% limit  166  is thus provided to a new switching time  168  at which the converter is switched “off”. The PWM control signals  170  for the second converter, illustrated in  FIG. 12 , may be similarly extended. Thus, PWM control signals to place the converter in a “on” state  172  may begin at the normal time in the overall cycle at which they would have otherwise begun, indicated by reference numeral  174 . However, a similar extension in the control may be made that is equal to that made for the first converter. Thus, the “on” period of the first converter, indicated by reference numeral  176  remains equal to that for the second converter, indicated by reference numeral  178 . Due to the magnetizing and demagnetizing dynamics of the transformers, a higher mean voltage output is obtained than would be available if the strict 50% duty cycle for each interleaved converter were held. 
     Implementation of the approach may follow the same logic as that described above. That is, the PWM duty cycle of the first converter may be determined based upon the process, power output requirements, and so forth. The actual “on” period for the first converter is then detected and recorded (e.g., by augmenting a counter based upon clock cycles during the “on” period), and a second converter is placed in the “on” state for an equal time. 
     Where processing capabilities or control logic does not permit such operation (e.g., where it is preferable to determine the full “on” period for the second converter prior to switching it to the “on” state), a delayed approach illustrated in  FIGS. 13 and 14  may be adopted. In this approach, the control signals  180  for the first converter will include a series of “on” PWM inputs  182 ,  184 ,  186 , and so forth. However, the control signals  188  for the second converter may be delayed one cycle to follow the first converter. That is, the control signals  188  for the second converter will include PWM commands  190 ,  192 ,  194 ,  196 , and so forth. The delay will result in a one-cycle shift of the PWM command of the first converter in its application to the second converter, as indicated for the “n−1” signals  184  and  194 , and the “n” commands  186  and  196  illustrated in both  FIGS. 13 and 14 . This delay allows the full duration of the “on” period of the second converter to be known before it is applied (in the next switching period). 
       FIGS. 15-19  are graphical representations of exemplary voltage and current waveforms of the type obtainable through use of the control scheme outlined in  FIGS. 11-14 . In particular,  FIG. 15  represents transformer primary control voltages  200  over time. As illustrated, when the converters are switched “on”, the voltage rises as indicated by blocks  202 , and when “off”, the voltage drops as indicated by reference numeral  204 . As shown in  FIG. 16 , then, the no-load magnetizing current  206 , then, rises, as indicated by reference numeral  208  when the transformer primary voltage is high, and declines, as indicated by reference numeral  210  when the voltage is low. As shown in  FIG. 17 , the full load primary current  212  rises, as indicated by reference numeral  214  during an initial period  216 . This initial rise corresponds to the leakage inductance current charge only. Then, as indicated by reference numeral  218 , the current rises at a lower rate during a period  220 , due to the effective secondary voltage and inductance. The decline in the full load primary current, as indicated by reference numeral  222 , occurs following removal of the primary voltage. 
     The full load secondary voltage is illustrated in  FIG. 18 . The voltage is shown as rising, as indicated by reference numerals  226  when the primary voltage is high, but with the “on” level shifted due to the leakage inductance of the transformer. When the primary voltage is removed, the secondary voltage falls, as indicated by reference numeral  228 . The full load magnetizing current  230  is illustrated in  FIG. 19 . Here, the current rises, as indicated by reference numeral  232  when the secondary voltage is high, and declines, as indicated by reference numeral  234  when the secondary voltage is low. An unutilized portion of the pulse width then results, as indicated by reference numeral  236 . The techniques described above allow for more full utilization of the pulse width during such periods by extending the command PWM signals beyond the 50% level, as discussed above. 
     Various modification and variations of the circuitry, systems and techniques described above may be envisaged. For example, while two inverters or converters have been described, the same techniques may be used in systems with more than two such circuits connected to provide common output. Also, while forward converters are illustrated and described, other converter types and applications may be used. Further, while digital, discrete techniques are described for determining switching of the converters, analog and hybrid circuitry may also be employed for this purpose. Finally, while equal time periods for switching of the converters are described, the same or similar techniques may be used for controlling loading (and heating) of the circuits by dissimilar switching periods (e.g., based on adding or subtracting from the counts of clock pulses of a first converter when controlling a second converter). 
     While in the foregoing discussion and examples two converters were utilized, it should be noted that the extended period control techniques discussed may be used in systems utilizing a single converter as well. While such converters (e.g., single ended converters) have typically been utilized with a 50% duty cycle limit, it has been determined that using the present teachings an extended duty cycle (i.e., not limited to a maximum of 50%) can be obtained. Here again, the present techniques may be used with systems having more than two converters as well. 
     It should also be noted that the PWM duty cycle implemented will typically be changed during operation of the system, and that the particularly duty cycle at any time may be dependent upon the load, such as the power drawn by the welding or cutting operation in the present context. Such control may be based on monitoring power and/or current drawn by the load. In one presently contemplated embodiment, for example, the PWM duty cycle might be extended by a desired number of clock pulses (or any other desired standard period) for each amp of output current drawn by the load. It may also be based on metering and/or determining a time between a point when the transformer primary current falls to zero and a start of a subsequent switching period, as shown in the foregoing graphical illustrations. 
     While only certain features of the invention have been illustrated and described herein, many modifications and changes will occur to those skilled in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the invention.