Patent Publication Number: US-8542571-B2

Title: Cell search method, forward link frame transmission method, apparatus using the same and forward link frame structure

Description:
CROSS-REFERENCE TO RELATED PATENT APPLICATION 
     This application claims the benefit of Korean Patent Application Nos. 10-2005-0107474 and 10-2006-0074308, respectively filed on Nov. 10, 2005 and Aug. 7, 2006, in the Korean Intellectual Property Office, the disclosures of which are incorporated herein by reference. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to an Orthogonal Frequency Division Multiplexing (OFDM) cellular system, and more particularly, to a method of searching for a cell in an OFDM cellular system, a mobile station using the method, a base station using the method, a system using the method, and a frame structure used in the method. 
     2. Description of the Related Art 
     In a Wideband Code Division Multiple Access (WCDMA) method of the 3 rd  Generation Partnership Project (3GPP), a cellular system uses a total of 512 long Pseudo-Noise (PN) scrambling codes in order to identify base stations of a forward link, wherein each adjacent base station uses a unique long PN scrambling code as a scrambling code of forward link channels. When a mobile station is turned on, the mobile station must acquire system timing of a base station to which the mobile station belongs (i.e., a base station of which a reception signal has the maximum amplitude) and a long PN scrambling code ID used by the base station. This process is called a mobile station&#39;s cell search process. 
     In WCDMA, in order to easily perform the mobile station&#39;s cell search process, the 512 long PN scrambling codes are grouped into 64 groups, and a primary sync channel and a secondary sync channel are included in the forward link. The primary sync channel is used for a mobile station to acquire slot sync, and the secondary sync channel is used for the mobile station to acquire a 10-msec frame boundary and long PN scrambling code group ID information. 
     The mobile station&#39;s cell search process in WCDMA is accomplished in 3 steps. In the first step, a mobile station acquires slot sync using a Primary Scrambling Code (PSC). In WCDMA, the same 15-slot PSC is transmitted every 10 msec, and PSCs transmitted by all base stations are the same signal. In the first step, slot sync is acquired using a matching filter suitable for the PSC. 
     In the second step, long PN scrambling code group ID information and a 10-msec frame boundary are acquired using the slot timing information acquired in the first step and a Secondary Scrambling Code (SSC). 
     In the third step, a long PN scrambling code ID used by a currently connected base station is acquired using the 10-msec frame boundary and the long PN scrambling code group ID information that were acquired in the second step and a common pilot channel code correlator. That is, since 8 long PN scrambling codes are mapped to a single code group, the mobile station detects the long PN scrambling code ID used in the current cell on the basis of outputs of the common pilot channel code correlator for 8 long PN scrambling codes corresponding to the long PN scrambling code group ID information. 
     In WCDMA, a sync channel consists of a primary sync channel and a secondary sync channel, and the primary sync channel, the secondary sync channel, a common pilot channel, and other data channels are multiplexed in a CDMA method based on a time domain direct sequence spread spectrum. 
     Recently, in the 3GPP, an OFDM-based wireless transmission technology standardization is being established as a part of 3 rd  Generation Long Term Evolution (3G-LTE) to compensate for disadvantages of WCDMA. The sync channel &amp; common pilot channel structure and the mobile station&#39;s cell search process used in WCDMA are suitable for Direct Sequence Code Division Multiple Access (DS-CDMA) but cannot be applied to an OFDM forward link. Thus, a forward link sync channel &amp; common pilot channel structure, a mobile station&#39;s initial cell search method, and an adjacent cell search method for handover are required in an OFDM cellular system. 
     SUMMARY OF THE INVENTION 
     The present invention provides a cell search apparatus and method in an Orthogonal Frequency-Division Multiplexing (OFDM) cellular system. 
     The present invention also provides a forward link frame transmission apparatus and method for supporting the cell search method. 
     The present invention also provides an OFDM cellular system to which the cell search method is applied. 
     The present invention also provides a computer readable recording medium storing a computer readable program for executing the cell search method. 
     The present invention also provides a structure of a forward link frame used in the cell search method. 
     According to an aspect of the present invention, there is provided a cell search apparatus in an OFDM cellular system in which a unique scrambling code is assigned to each cell, the cell search apparatus including: a sync acquirer acquiring synchronization of sync channel symbols using a sync channel of a forward link; and a group detector detecting at least one hopping codeword element belonging to a hopping codeword of a target cell from a forward link signal containing sync channel symbols sequence-hopped using a hopping codeword corresponding to a code group to which a scrambling code of each cell belongs based on the acquired synchronization, and detecting a code group of the target cell based on the detected hopping codeword element, wherein the hopping codewords are orthogonal to a cyclic shift operation. 
     According to another aspect of the present invention, there is provided a cell search apparatus in an OFDM cellular system in which a unique scrambling code is assigned to each cell, the cell search apparatus including: a sync acquirer acquiring synchronization of sync channel symbols using a sync channel of a forward link; and a boundary detector detecting at least one hopping codeword element belonging to a hopping codeword of a target cell from a forward link signal containing sync channel symbols sequence-hopped using a hopping codeword that is orthogonal to a cyclic shift operation for each cell based on the acquired synchronization and detecting a frame boundary of the target cell based on the detected hopping codeword element. 
     According to another aspect of the present invention, there is provided a forward link frame transmission apparatus of a base station belonging to an OFDM cellular system in which a unique scrambling code is assigned to each cell, the forward link frame transmission apparatus including: a sync channel generator generating sync channel sequences corresponding to elements of a hopping codeword corresponding to a code group to which a scrambling code of a base station belongs; and a frame transmitter performing sequence hopping of each sync channel symbol using each generated sync channel sequence, generating an OFDM symbol-based forward link frame containing the sequence-hopped sync channel symbols, and transmitting the generated forward link frame, wherein hopping codewords used in the system are orthogonal to cyclic shift. 
     According to another aspect of the present invention, there is provided an OFDM cellular system including a mobile station and a plurality of base stations, and in which a unique scrambling code is assigned to each cell, each of the plurality of base stations including: a sync channel generator generating a hopping codeword corresponding to a code group to which a scrambling code of the base station belongs and generating sync channel sequences corresponding to elements of the generated hopping codeword; and a frame transmitter performing sequence hopping of each sync channel symbol using each generated sync channel sequence, generating an OFDM symbol-based forward link frame containing the sequence-hopped sync channel symbols, and transmitting the generated forward link frame, and the mobile station including: a sync acquirer acquiring synchronization of sync channel symbols using a sync channel of a forward link; and a group detector detecting at least one hopping codeword element belonging to a hopping codeword of a target cell from a forward link signal containing sync channel symbols sequence-hopped using a hopping codeword of each cell based on the acquired synchronization and detecting a code group of the target cell based on the detected hopping codeword, wherein the hopping codewords are orthogonal to a cyclic shift operation. 
     According to another aspect of the present invention, there is provided a cell search method in an OFDM cellular system in which a unique scrambling code is assigned to each cell, the cell search method including: (a) acquiring synchronization sync channel symbols using a sync channel of a forward link; and (b) detecting at least one hopping codeword element belonging to a hopping codeword of a target cell from a forward link signal containing sync channel symbols sequence-hopped using a hopping codeword corresponding to a code group to which a scrambling code of each cell belongs based on the acquired synchronization, and detecting a code group of the target cell based on the detected hopping codeword element, wherein the hopping codewords are orthogonal to a cyclic shift operation. 
     According to another aspect of the present invention, there is provided a cell search method in an OFDM cellular system in which a unique scrambling code is assigned to each cell, the cell search method including: (a) acquiring synchronization of sync channel symbols using a sync channel of a forward link; and (b) detecting at least one hopping codeword element belonging to a hopping codeword of a target cell from a forward link signal containing sync channel symbols sequence-hopped using a hopping codeword that is orthogonal to a cyclic shift operation for each cell based on the acquired synchronization, and detecting a frame boundary of the target cell based on the detected hopping codeword element. 
     According to another aspect of the present invention, there is provided a forward link frame transmission method of a base station belonging to an OFDM cellular system in which a unique scrambling code is assigned to each cell, the forward link frame transmission method including: (a) generating a hopping codeword corresponding to a code group to which a scrambling code of the base station belongs, and generating sync channel sequences corresponding to elements of the generated hopping codeword; and (b) performing sequence hopping of each sync channel symbol using each generated sync channel sequence, generating an OFDM symbol-based forward link frame containing the sequence-hopped sync channel symbols, and transmitting the generated forward link frame, wherein the hopping codewords are orthogonal to a cyclic shift operation. 
     According to another aspect of the present invention, there is provided a structure of a forward link frame in an OFDM cellular system in which a unique scrambling code is assigned to each cell, the forward link frame including sync channel symbols sequence-hopped using sync channel sequences corresponding to elements of a hopping codeword corresponding to a code group to which a scrambling code belongs, wherein the hopping codewords are orthogonal to a cyclic shift operation. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other features and advantages of the present invention will become more apparent by describing in detail exemplary embodiments thereof with reference to the attached drawings in which: 
         FIG. 1  illustrates a structure of a forward link frame according to an embodiment of the present invention; 
         FIG. 2  is a conceptual diagram for explaining how to group scrambling codes according to an embodiment of the present invention; 
         FIG. 3  illustrates a sub-frame containing a sync channel symbol according to an embodiment of the present invention; 
         FIG. 4  illustrates a structure of a sync channel symbol in the time domain according to an embodiment of the present invention; 
         FIG. 5  is a block diagram of a base station according to an embodiment of the present invention; 
         FIGS. 6A and 6B  are a block diagram and a conceptual diagram, respectively, of a diversity controller in a case where delay diversity is applied to the base station illustrated in  FIG. 5 , according to an embodiment of the present invention; 
         FIG. 7  is a block diagram of a receiver of a mobile station according to an embodiment of the present invention; 
         FIG. 8  is a block diagram of a sync acquirer of the receiver illustrated in  FIG. 7 , according to an embodiment of the present invention; 
         FIG. 9  is a graph illustrating differential correlation values calculated by a differential correlator illustrated in  FIG. 8  based on sample positions according to an embodiment of the present invention; 
         FIG. 10  illustrates a structure of an input signal provided to a group detector illustrated in  FIG. 7  based on a sync channel Orthogonal Frequency-Division Multiplexing (OFDM) symbol timing acquired by the sync acquirer illustrated in  FIG. 7 , according to an embodiment of the present invention; 
         FIG. 11  is a block diagram of the group detector illustrated in  FIG. 7 , according to an embodiment of the present invention; 
         FIG. 12  is a block diagram of a group &amp; boundary detector illustrated in  FIG. 11 , according to an embodiment of the present invention; 
         FIG. 13  is a graph illustrating outputs of a code correlation calculator illustrated in  FIG. 12 , according to an embodiment of the present invention; 
         FIG. 14  illustrates correlation values stored in a correlation buffer illustrated in  FIG. 12 , according to an embodiment of the present invention; 
         FIG. 15  illustrates a structure of frame information acquired in a second step of a cell search process according to an embodiment of the present invention; 
         FIG. 16  is a block diagram of a code detector illustrated in  FIG. 7 , according to an embodiment of the present invention; 
         FIG. 17  is a conceptual diagram for explaining an operation of a pilot correlator according to an embodiment of the present invention; 
         FIG. 18  is a block diagram of the sync acquirer of the receiver illustrated in  FIG. 7 , according to another embodiment of the present invention; 
         FIG. 19  is a conceptual diagram for explaining an operation of a frequency offset switching unit according to an embodiment of the present invention; 
         FIG. 20  is a flowchart illustrating a cell search method of a mobile station according to an embodiment of the present invention; and 
         FIG. 21  is a flowchart illustrating a forward link frame transmission method of a base station according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention will now be described more fully with reference to the accompanying drawings, in which exemplary embodiments of the invention are shown. 
     In general, each base station of an Orthogonal Frequency Division Multiplexing (OFDM) cellular system scrambles OFDM symbols using a long PN scrambling code. However, since the base station can use another scrambling code instead of the long PN scrambling code, any code used to scramble OFDM symbols is hereinafter called a scrambling code for convenience of description. 
     Although each base station according to an embodiment of the present invention can achieve transmission diversity using a method of including a plurality of transmission antennas, a delay diversity method, or other similar methods, it is assumed in the present specification for convenience of description that each of the base stations includes 2 transmission antennas. 
     Though a mobile station according to an embodiment of the present invention can achieve reception diversity using a method of including a plurality of reception antennas or other similar methods, it is assumed in the present specification for convenience of description that the mobile station includes 2 reception antennas. This mobile station must combine data of data paths according to the reception diversity. Though a simple summing method is used in the present specification as a data combining method, it will be understood by those of ordinary skill in the art that the data combining method is not limited to the simple summing method. 
     The present invention relates to a method of performing a cell search including sync acquisition, frame boundary detection, and scrambling code detection. 
     The term ‘sync acquisition’ will be used in the present specification as a comprehensive term for sync channel symbol timing detection, sync block timing detection, and sync block boundary detection. That is, since a position of a sync channel symbol can be obtained by detecting sync block timing, sync channel symbol timing is equivalent to the sync block timing. The term ‘sync information’ will be used in the present specification as a comprehensive term for information on sync channel symbol timing, information on sync block timing, and information on a sync block boundary. The term ‘frame boundary detection’ will be used in the present specification as a comprehensive term for frame boundary timing detection. The term ‘frame boundary information’ will be used in the present specification as a comprehensive term for information on frame boundary timing. 
     The term ‘code group detection’ will be used in the present specification as a comprehensive term for code group identifier detection and code group detection, and the term ‘code group information’ will be used in the present specification as a comprehensive term for a code group identifier and a code group. The term ‘scrambling code detection’ will be used in the present specification as a comprehensive term for scrambling code identifier detection and scrambling code detection, and the term ‘scrambling code information’ will be used in the present specification as a comprehensive term for a scrambling code identifier and a scrambling code. 
     A sync channel sequence is a sequence mapped to subcarriers occupied by sync channel in a frequency domain of a sync channel symbol, each element of the sync channel sequence being used as a Fourier coefficient in an occupied subcarrier frequency. In the present invention, a sequence indicates the sync channel sequence, and the number of sync channel sequences used in a system is the same as a code alphabet size of a hopping code used in the system. 
     A sync channel symbol signal (time domain signal) transmitted from every sync channel symbol position is generated by an inverse Fourier transform of the sync channel sequence, a different sync channel sequence can be used at every sync channel position, and a sequence index of a sync channel sequence used at every sync channel symbol position is an element of a hopping codeword assigned to a relevant base station. 
     A set of hopping codewords used in the system is defined as a hopping code. 
     The term ‘Fourier transform’ will be used for convenience of description in the present specification as a comprehensive term for discrete Fourier transform and fast Fourier transform. 
       FIG. 1  illustrates a structure of a forward link frame according to an embodiment of the present invention. 
     Referring to  FIG. 1 , the forward link frame has a 10-msec duration and includes 20 sub-frames  110 . In  FIG. 1 , the horizontal axis represents time, and the vertical axis represents frequency (OFDM subcarrier). Each of the 20 sub-frames  110  has a 0.5-msec length and includes 7 or 6 OFDM symbols  120  including at least one common pilot symbol  130 . In addition, each of the 20 sub-frames  110  includes a single or no sync channel symbol  100 . In the current embodiment, a single sync channel symbol duration exists at every 5 sub-frames  110 , and thus a total of 4 sync channel symbol durations exist in the forward link frame. In this case, a sync channel symbol repetition period  140  is the same as a length obtained by summing lengths of 5 sub-frames  110 , and thus a repetition period of the total sync channel symbols in the forward link frame is 4. For convenience of description, the sync channel symbol repetition period  140  is called a sync block. That is,  FIG. 1  shows that the number N b  of sync blocks in a single frame (10 msec) is 4. 
     The OFDM symbols  120  that remain due to the exclusion of the sync channel symbols  100  are multiplied by a cell-specific scrambling code in a frequency domain in order to identify the cell. In the forward link frame illustrated in  FIG. 1 , a scrambling code having a scrambling code identifier (ID) of 8 and belonging to a code group having a code group ID of 0 is used. The scrambling code will be described later in detail. 
     The forward link frame according to an embodiment of the present invention has a structure whereby sync channel sequences, which are indicated by respective elements of a hopping codeword assigned to a base station, are assigned to respective sync channel symbols. In the present specification, a method of transmitting a sync channel sequence indicated by each element of a hopping codeword at every sync block is defined as sequence hopping of sync channel, wherein each hopping pattern, i.e., each hopping codeword, respectively corresponds to each scrambling code group. In  FIG. 1 , the used hopping codeword is (2, 5, 8, 11). 
     When a mobile station is initially turned on, the mobile station must first detect a 10-msec frame boundary  150  of a current cell to which the mobile station belongs and detect a scrambling code used by the current cell. The scrambling code detection is performed in order to detect the scrambling code ID  8  contained in the forward link frame illustrated in  FIG. 1 , which is a scrambling code ID of the current cell. This is called a mobile station&#39;s cell search process. 
     According to an OFDM cellular system according to an embodiment of the present invention, cell-specific scrambling codes are grouped into a plurality of code groups, each code group containing at least one scrambling code. Thereafter, information on a code group to which a scrambling code of a current cell belongs and information on a frame boundary are inserted into a hopping codeword assigned to sync channel symbols. That is, each hopping codeword specifies a code group and a frame boundary of each cell. 
     A mobile station can perform the cell search process using a forward link frame containing sync channel symbols, which are generated by performing the above-described process, and common pilot channel symbols. Since a mobile station can simultaneously detect a frame boundary and a code group using a sync channel having a single code structure, the mobile station can efficiently perform the cell search process. In addition, since the mobile station detects only a scrambling code belonging to the detected code group, complexity in a code detection process can be reduced. Pilot correlation is used in a code detection method, which will be described later in detail. 
       FIG. 2  is a conceptual diagram for explaining how to group scrambling codes according to an embodiment of the present invention. 
     A scrambling code or scrambling code ID  200  used to scramble common pilot symbols or data symbols is assigned to each base station belonging to an OFDM cellular system. In particular, the OFDM cellular system according to an embodiment of the present invention groups the scrambling codes into code groups. That is, at least one scrambling code ID exists in each code group. Referring to  FIG. 2 , the number of scrambling codes used in the OFDM cellular system is at least 18, and the number of code groups is at least 4. Thus, each of the 4 code groups contains 8 scrambling code IDs  200  or scrambling codes. In particular, if each code group contains only one scrambling code, code groups respectively correspond to scrambling codes, and thus a hopping codeword can specify not only a code group but also a scrambling code. 
       FIG. 3  illustrates a sub-frame containing a sync channel symbol according to an embodiment of the present invention, e.g., a first sub-block  110  of a first sync block as illustrated in  FIG. 1 . 
     According to the sub-frame illustrated in  FIG. 3 , a first OFDM symbol  100  includes data symbol  330  and sync symbol  320  which is an element of a sync channel sequence assigned to the sync channel symbol, and a second OFDM symbol  130  includes pilot symbol  340  and the data symbol  330 . As described above, the first OFDM symbol  100  is a sync channel symbol, and the second OFDM symbol  130  is a common pilot channel symbol. The sub-frame illustrated in  FIG. 3  is just an illustration, and the sync channel symbol can be placed at another OFDM symbol in the sub-frame. The important feature is that a sync channel symbol position in every sync block is the same. That is, each interval between adjacent sync channels is the same. However, when symbols per one sub-frame are 6 and 7, the lengths of a cyclic prefix (CP) are different. Accordingly, when the numbers of the OFDM symbols in the sub-frames are 6 and 7, the sync channel symbol may be placed at the end of the sub-frames. 
     A sync channel sequence which is each element of a hopping codeword is assigned to each sync channel symbol, and each element of the assigned sync channel sequence is carried on each subcarrier belonging to a sync channel occupied bandwidth. As a method of assigning the sync channel occupied bandwidth, a sync channel can occupy a band, which remains, by excluding a guard band or by occupying a portion of the remaining band. An example of a system to which the latter method can be applied is a system which must support a scalable bandwidth, such as a 3 rd  Generation Long Term Evolution (3G-LTE) system. That is, a mobile station using only 1.25 MHz, a mobile station using only 2.5 MHz, and all mobile stations using 5 MHz, 10 MHz, 15 MHz, and 20 MHz can acquire sync with a base station system when sync channel occupies only a portion of a total system bandwidth  310  as illustrated in  FIG. 3 . For example, when the system bandwidth  310  is 10 MHz, only 1.25 MHz in the center, which remains due to the exclusion of a DC subcarrier, is used. The 3G LTE system supports the bandwidth having a minimum standard of 10 MHz, and so when the system bandwidth is 20 MHz, the system bandwidth may have 1.25 MHz sync channel bands at each side of the 10 MHz bands in order for the mobile station to easily search adjacent cells during communication. 
     A cell search unit of a mobile station, which will be described later, can increase cell search performance by performing filtering so as to pass only a sync channel occupied band  300  illustrated in  FIG. 3 . 
     Referring to  FIG. 3 , a sync channel using one of 7 OFDM symbol durations in a sub-frame occupies a partial band  300  out of the entire band  310  as described above. The sync channel can use all subcarriers in the sync channel occupied band  300  or only one of 2 adjacent subcarriers as illustrated in  FIG. 3 . A predetermined value, e.g. 0, is assigned to the unused subcarrier in the latter method. In particular, if the latter method is used, a time domain signal of a sync channel symbol excluding a cyclic prefix has a pattern repeated in a time domain, which will be described later with reference to  FIG. 4 . 
     In  FIG. 3 , C (k) =[c 0   (k) , c 1   (k) , c 2   (k) , . . . , c N-1   (k) ] indicates a sync channel sequence in which a hopping codeword element corresponding to a relevant sync channel symbol is k. Each element of the sync channel sequence, i.e., c 0   (k) , c 1   (k) , c 2   (k) , . . . , c N-1   (k) , has a value of a complex number, and is transmitted by being assigned to a subcarrier  320  belonging to the sync channel occupied band  300  illustrated in  FIG. 3 . An arbitrary sequence can be used as the sync channel sequence. For example, a Generalized Chirp Like (GCL) sequence defined using Equation 1 can be used as the sync channel sequence. 
     
       
         
           
             
               
                 
                   
                     
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     In Equation 1, k is defined as an index of an arbitrary element of a hopping codeword and denotes a sync channel sequence number, C n   (k)  denotes an n th  element of a k th  sync channel sequence, N denotes the length of the GCL sequence. In particular, in the GCL sequence, each code length N is a prime number, and a total of N−1 sequences exist. That is, if the GCL sequence is used, a GCL sequence set used in a system consists of N−1 GCL sequences. The number of GCL sequences is the same as a code alphabet size of a hopping code. 
     A hopping codeword assigned to each base station is transmitted to a mobile station in the form of sync channel sequence hopping of a forward link frame. That is, in an embodiment of the present invention, the base station maps hopping codeword elements of which is a GCL sequence index to respective sync channel symbols, and transmits the sync channel symbols to a mobile station, which allows the mobile station to detect a hopping codeword included in the sync channel symbols transmitted from a target base station. Here, an example of the target base station can be a base station for which the mobile station initially searches or an adjacent base station to be searched so as to allow handover to occur. 
     The common pilot channel symbol  130  uses one or two of 7 or 6 OFDM symbol durations in the sub-frame illustrated in  FIG. 3 , and in the common pilot channel symbol duration, the pilot symbol  340  and the data symbol  330  can be multiplexed by using a Frequency Division Multiplexing (FDM) method. The common pilot channel is used to estimate a channel for coherent decoding of a data channel of a forward link and detect a scrambling code or a scrambling code ID in a third step of the cell search process according to an embodiment of the present invention, which will be described later in detail. 
     Table 1 is a table illustrating a set of sync channel hopping patterns of code groups, i.e., a set of hopping codewords, in a case where the number of code groups is 3 and the number of sync channel symbols in a forward link frame is 4 as illustrated in  FIG. 2 . That is, the 3 hopping patterns can be represented using hopping codewords, each having a length of 4, and the length of each hopping codeword is the same as the number of sync channel symbols per 10-msec frame. An entire set of hopping codewords is defined as a hopping code. Each base station uses the same sync channel hopping pattern (hopping codeword) for every frame, and base stations having different group numbers use different hopping codewords. Referring to Table 1, each hopping codeword consists of 4 hopping codeword elements and each hopping codeword respectively corresponds to a code group ID. 
     
       
         
           
               
               
               
             
               
                   
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                 3, 6, 9, 12 
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     In Table 1, an alphabet size of a hopping code sequence is 40. That is, a hopping codeword element k transmitted through each sync block is one of numbers 1 to 40. For example, if a long PN scrambling code ID of a current base station is 8, the current base station belongs to a code group No. 2 (referring to  FIG. 2 ), and a hopping codeword assigned to the code group No. 2 is {2, 5, 8, 11} (referring to Table 1). Thus, 4 sync channel symbols transmitted through each frame by the current base station respectively have hopping codeword elements of 2, 5, 8, and 11, and values defined by Equation 1 according to the hopping codeword element k are assigned to a subcarrier used by each of the 4 sync channel symbols. In particular,  FIG. 1  shows an illustration of a case where a code group ID of the current base station is 2. 
     Specifically in Table 1, the number of scrambling codes is 12, which is equal to the number of code groups multiplied by the number of sync channel symbols in the frames. This is a condition ensuring the hopping patterns are orthogonal to all the cyclic shifts. Such a condition allows the mobile station to acknowledge a code group ID and a frame boundary to which the scrambling code belongs, and the scrambling code used by the current base station, when the mobile station demodulates only one sync channel symbol. 
     Meanwhile, a method of dividing one code sequence having a very long sequence length as much as the number of the sync channel symbol and allotting each divided code sequence to the sync channel symbol is also a kind of sequence hop, and one of ordinary skill in the art can easily understand that the method is in the range of the present invention using hopping codewords having orthogonality. An example of applying the GCL sequence, defined by Equation 1, as the code sequence of the above method is as follows. In this case, the number of sync channel symbols in one frame is 4 as shown in  FIG. 1 , and the number of sequence elements in one the sync channel symbol is 40. Also, in this case, a cell belonging to a code group 1 uses a GCL sequence corresponding to N=161 and k=1 as the code sequence, and a cell belonging to a code group 2 uses a GCL sequence corresponding to N=161 and k=2 as the code sequence. That is, the cell belonging to the code group 1 divides the code sequence into 4 sync channel sequences, and sequence-hops each sync channel symbol using the sync channel sequences. Here, an example of dividing the code sequence includes using sequence elements corresponding to n=0, 1, through to 39 from among the sequence elements of the code sequence in a first sync channel symbol and using sequence elements corresponding to n=40, 41 through to 79 in a second sync channel symbol, but is not limited thereto. 
       FIG. 4  illustrates a structure of a sync channel symbol in the time domain according to an embodiment of the present invention. 
     Referring to  FIG. 4 , N T  denotes the number of samples of the entire sync channel OFDM symbol duration, N CP  denotes the number of samples of a cyclic prefix (CP) duration  350 , and N S  denotes the number of samples of a symbol duration  370  excluding the CP duration  350 . In particular, if the sync channel symbol uses only odd-th or even-th subcarriers in a sync channel occupied band for transmission of a relevant sync channel sequence and the sync channel symbol allocates a predetermined value (e.g., 0) to remaining subcarriers, a first duration  360  and a second duration  365  forming the duration denoted by reference numeral  370  have a specific pattern. If the sync channel symbol uses DC component subcarriers, the first duration  360  and the second duration  365  have the same waveform in a time domain signal of a transmitter end, and if the sync channel symbol does not use the DC component subcarriers, the second duration  365  has a waveform 180° phase reversed from a waveform of the first duration  360 . By using this time domain repetition pattern of the sync channel symbol, sync can be acquired with a simple structure using a differential correlation operation, which will be described later in detail. The first duration  360  and the second duration  365  may also be symmetrical to each other. In this case, reverse differential correlation can be used. The differential correlation and the reverse differential correlation used in a sync acquisition process are within the spirit and scope of the present invention. 
     A base station according to an embodiment of the present invention transmits information contained in a hopping codeword assigned thereto to a mobile station in the base station&#39;s cell by carrying the information on a sync channel as illustrated in  FIG. 1 . That is, the code group information and 10-msec frame boundary information are contained in the hopping codeword and transmitted to the mobile station. 
     Thus, the sync channel allows the mobile station to achieve sync block timing in a first step of a cell search process and acquire a 10-msec frame timing and code group information in a second step of the cell search process. That is, according to an embodiment of the present invention, using only a single sync channel, the sync block timing is achieved_in the first step of the cell search process and the 10-msec frame timing and the code group information are acquired in the second step of the cell search process. 
     A primary sync channel is used in the first step of the cell search process and a secondary sync channel is used in the second step of the cell search process in a conventional WCDMA system, whereas the same single sync channel is used in the first and second steps of the cell search process according to an embodiment of the present invention. In addition, a sync channel used in the conventional WCDMA system is a signal spread in a time domain direct sequence spread spectrum, whereas a sync channel used in an embodiment of the present invention is a signal transmitted by being scrambled using a frequency domain code and OFDM modulation. 
       FIG. 5  is a block diagram of a base station according to an embodiment of the present invention. Referring to  FIG. 5 , the base station includes a sync channel generator  400 , a common pilot channel generator  401 , a data channel generator  402 , a diversity controller  403 , OFDM symbol mappers  404 -A and  404 -B, scramblers  405 -A and  405 -B, inverse Fourier transformers  406 -A and  406 -B, CP insertion units  407 -A and  407 -B, intermediate frequency/radio frequency (IF/RF) units  408 -A and  408 -B, and transmission antennas  409 -A and  409 -B. 
     The data channel generator  402  generates data symbol that is to be transmitted such as reference numeral  330  of  FIG. 3 , and the common pilot channel generator  401  generates pilot symbol  340  illustrated in  FIG. 3 . The sync channel generator  400  generates a sync channel sequence, such as the sync channel sequence  320  illustrated in  FIG. 3 , corresponding to each element of a hopping codeword assigned to the base station. That is, if the hopping codeword assigned to the base station is {2, 5, 8, 11}, the sync channel generator  400  generates a sequence, i.e. N elements, obtained by substituting k=2 into Equation 1 for a first sync block. If the number of frequency domain subcarriers which a sync channel symbol can use for transmission of a relevent sync channel sequence is carried is less than N, e.g., if N=41 and the number of subcarriers occupied by the sync channel sequence is 38, the last 3 elements of the sync channel sequence defined using Equation 1 are not transmitted. 
     Each of the OFDM symbol mappers  404 -A and  404 -B maps data values of the data channel, the pilot channel, and the sync channel to positions in the frequency domain as illustrated in  FIG. 3 . Each of the scramblers  405 -A and  405 -B multiplies an output of each of the OFDM symbol mappers  404 -A and  404 -B, i.e., OFDM symbols excluding a sync channel symbol from the mapping result, by a base station&#39;s unique scrambling code in the frequency domain. 
     Each of the inverse Fourier transformers  406 -A and  406 -B generates a time domain signal by performing an inverse Fourier transform on the output of each of the scramblers  405 -A and  405 -B Each of the CP insertion units  407 -A and  407 -B inserts a CP for enabling demodulation of an OFDM signal, even with a channel multi-path delay, into the output of each of the inverse Fourier transformers  406 -A and  406 -B. Each of the IF/RF units  408 -A and  408 -B up-converts an output signal of each of the CP insertion units  407 -A and  407 -B, which is a baseband signal, to a band pass signal and amplifies the up-converted signal. 
     Each of the transmission antennas  409 -A and  409 -B transmits the amplified signal. 
     In  FIG. 5 , the number of transmission antennas  409 -A and  409 -B is 2. That is, if the base station according to an embodiment of the present invention has only one transmission antenna  409 -A without the transmission antenna  409 -B, the OFDM symbol mapper  404 -B, the scrambler  405 -B, the inverse Fourier transformer  406 -B, the CP insertion unit  407 -B, the IF/RF unit  408 -B, and the diversity controller  403  can be omitted. 
       FIG. 5  illustrates a case where sync channel symbols are transmitted while achieving transmission diversity using 2 transmission antennas at a transmitter end of the base station. The transmission diversity using the diversity controller  403  illustrated in  FIG. 5  will now be described. Sync channel symbols belonging to adjacent sync blocks are transmitted through different transmission antennas in order to achieve spatial diversity. For example, a sync channel symbol belonging to a first sync block is transmitted through the first transmission antenna  409 -A, a sync channel symbol belonging to a second sync block is transmitted through the second transmission antenna  409 -B, and a sync channel symbol belonging to a third sync block is transmitted through the first transmission antenna  409 -A. This switching so as to achieve the spatial diversity is performed by the diversity controller  403 . That is, using a method of applying Time Switching Transmit Diversity (TSTD) to the sync channel, the diversity controller  403  provides an output of the sync channel generator  400  to the OFDM symbol mapper  404 -A or  404 -B by switching the output of the sync channel generator  400 . 
     Besides the spatial diversity or the TSTD diversity, delay diversity can be used as the transmission diversity. 
       FIGS. 6A and 6B  are a block diagram and a conceptual diagram, respectively, of the diversity controller  403  in a case where the delay diversity is applied to the base station illustrated in  FIG. 5 , according to an embodiment of the present invention. 
     Referring to  FIG. 6A , the diversity controller  403  includes a delay weight multiplier  410 . N sync symbols as illustrated in  FIG. 3  are generated at every sync channel symbol by the sync channel generator  400  illustrated in  FIG. 5 . The generated sync symbols are separated into two data paths. According to the upper data path, the sync symbols are directly provided to the OFDM symbol mapper  404 -A. According to the lower data path, the sync symbols are input to the delay weight multiplier  410 , and the output of the delay weight multiplier  410  is input to the OFDM symbol mapper  404 -B. 
       FIG. 6B  is a conceptual diagram for explaining an operation of the delay weight multiplier  410 . 
     Referring to  FIG. 6B , the delay weight multiplier  410  delays the generated sync symbols and includes N multipliers. 
     Each of the N multipliers multiplies each sync symbol assigned to each subcarrier used by a sync channel symbol, i.e., each of the N pieces of generated sync symbol, by a weight. A weight w(n) multiplied by sync symbol assigned to an n th  subcarrier used by the sync channel symbol is calculated using Equation 2.
 
 w ( n )=exp(− j 2π n· 2 D   m   /N   s ),  n= 0, 1, 2, . . . ,  N− 1  (2)
 
     In Equation 2, D m  denotes a delay of an FFT sample unit in the time domain for an m th  transmission antenna, and N s  denotes the number of FFT samples. Since it is assumed, as illustrated in  FIG. 3 , that sync symbol is carried on every other subcarrier, 2D m  is used instead of D m . If the number of transmission antennas  409 -A and  409 -B is 2 as illustrated in  FIG. 5 , a delay to the first transmission antenna  409 -A is D 0 =0, and a delay to the second transmission antenna  409 -B is D 1 . 
     The mobile station&#39;s cell search process according to an embodiment of the present invention is accomplished in 3 steps. The first step is a sync acquisition step, the second step is a step of detecting a code group and a frame boundary based on a sync block boundary acquired in the first step, and the third step is a step of detecting a scrambling code ID or a scrambling code of a current cell to which a mobile station belongs using the frame boundary and code group information acquired in the second step. In particular, in the second step of the cell search process, frequency offset estimation can be further included in order to increase a cell search&#39;s probability of success. In addition, after the third step of the cell search process, a fine timing/fine frequency detection step can be further performed. 
       FIG. 7  is a block diagram of a receiver of a mobile station according to an embodiment of the present invention. The mobile station has at least one reception antenna, and  FIG. 7  illustrates a case where the mobile station has 2 reception antennas. 
     Referring to  FIG. 7 , the receiver of the mobile station includes reception antennas  500 -A and  500 -B, down-converters  510 -A and  510 -B, a cell search unit  600 , a data channel demodulator  520 , a controller  530 , and a clock generator  540 . 
     RF signal type frames transmitted from base stations are received through the reception antennas  500 -A and  500 -B and converted to baseband signals S 1  and S 2  by the down-converters  510 -A and  510 -B. 
     The cell search unit  600  searches for a target cell using a sync channel symbol and a common pilot channel symbol included in the down-converted signals S 1  and S 2 . As a result of the cell search, sync channel symbol timing, a frame boundary, and a long PN scrambling code of the target cell can be detected, and the target cell is, for example, searched for when the mobile station searches an initial cell at the first time or an adjacent cell, so as to allow handover to occur. 
     The controller  530  controls the cell search unit  600  and the data channel demodulator  520 . That is, the controller  530  controls timing and descrambling of the data channel demodulator  520  based on a cell search result acquired by controlling the cell search unit  600 . The data channel demodulator  520  demodulates data channel, such as data symbol  330  illustrated in  FIG. 3 , included in the down-converted signals S 1  and S 2  under control of the controller  530 . All the hardware in the mobile station operates by being synchronized with a clock generated by the clock generator  540 . 
     The cell search unit  600  includes sync channel band filters  610 -A and  610 -B, a sync acquirer  620 , a group detector  640 , and a code detector  680 . 
     The sync channel band filters  610 -A and  610 -B perform band pass filtering in order to pass only the sync channel occupied band  300  from among the entire OFDM signal band  310  illustrated in  FIG. 3 , with respect to the down-converted signals S 1  and S 2 . 
     The sync acquirer  620  acquires sync information S 5  using a sync channel symbol included in the filtered signals S 3  and S 4 . 
     The group detector  640  detects code group information S 7  and 10-msec frame boundary information S 6  using the acquired sync information S 5  and the 3 codewords illustrated in Table 1 pre-stored in a memory (not shown) of the mobile station. The group detector  640  can increase detection performance by performing frequency offset estimation and compensation before detecting the code group information S 7  and the 10-msec frame boundary information S 6 . In this case, an estimated frequency offset value can be provided to the code detector  680 . 
     The code detector  680  detects a scrambling code by means of a pilot correlation of a common pilot channel symbol included in the down-converted signals S 1  and S 2  based on the detected code group information S 7  and 10-msec frame boundary information S 6 . Although the common pilot channel symbol is extracted from the down-converted signals S 1  and S 2  as illustrated in  FIG. 7 , if the common pilot channel symbol is not influenced by the sync channel band filters  610 -A and  610 -B, the common pilot channel symbol can be extracted from the outputs S 3  and S 4  of the sync channel band filters  610 -A and  610 -B. In detail, the code detector  680  extracts the common pilot channel symbol by obtaining a position of the common pilot channel symbol based on the detected 10-msec frame boundary information S 6 , calculates correlation values between the extracted common pilot channel symbol and scrambling codes corresponding to scrambling code IDs belonging to the detected code group S 7  selected from among pre-stored scrambling code IDs, and detects a scrambling code ID used by a current base station based on the correlation calculation result. 
       FIG. 8  is a block diagram of the sync acquirer  620  of the receiver illustrated in  FIG. 7 , according to an embodiment of the present invention. Referring to  FIG. 8 , the sync acquirer  620  includes differential correlators  621 -A and  621 -B, an accumulator  623 , and a timing determiner  624 . In  FIG. 8 , it is assumed that sync channel symbols use even-th or odd-th subcarriers from among subcarriers belonging to the sync channel occupied band  300  illustrated in  FIG. 3 . 
     Each of the differential correlators  621 -A and  621 -B multiplies a sample value of each of the output signals S 3  and S 4  of the sync channel band filters  610 -A and  610 -B by a sample value received previously to the current sample value by a time corresponding to an N S /2 sample using the time domain signal repetition characteristic of sync channel OFDM symbols illustrated in  FIG. 3  and accumulates the multiplication result. Here, N S /2 denotes half of the OFDM symbol duration  370  excluding the CP duration  350  as illustrated in  FIG. 3  and corresponds to the first duration  360  or the second duration  365 . 
     Equations 3 and 4 represent outputs of the differential correlators  621 -A and  621 -B at an arbitrary sample point n according to an embodiment of the present invention. 
     
       
         
           
             
               
                 
                   
                     
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     In Equations 3 and 4, ( )* denotes a complex conjugate value, a denotes a reception antenna index having 0 or 1, and r a (n) denotes a sample value of a signal received at an n th  sample point through an a th  reception antenna. 
     A square of an absolute value and the absolute value are obtained in Equations 3 and 4 in order to maintain performance of the sync acquirer  620  regardless of an initial frequency offset. If the absolute value is not obtained in Equation 3 or 4, the performance of the sync acquirer  620  may be decreased in a state where the initial frequency offset is large. 
     The output of each of the differential correlators  621 -A and  621 -B, which is represented by Equation 3 or 4, is generated having a length of 5×7×N T  per sync block with reference to  FIG. 1 , and the timing determiner  624  detects a position of a sample, which generates a peak value, among these differential correlation values and determines the detected sample position as sync channel symbol timing. However, the sync acquirer  620  according to an embodiment of the present invention may include the accumulator  623  in order to increase symbol sync detection performance. 
     The accumulator  623  combines the outputs of the differential correlators  621 -A and  621 -B, which are represented by Equation 3 or 4 with respect to two antennas, calculates antenna combining values at 5×7×N T  sample positions, and accumulates each antenna combining value for samples separated by every sync block length from each sample position. That is, an output γ(n) of the accumulator  623  can be represented by Equation 5. 
     
       
         
           
             
               
                 
                   
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                     ⁡ 
                     
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     Here, z(n) denotes a sum of differential correlation values of the reception antennas represented by Equation 3 or 4, L denotes a sync block length, i.e. 5×7×N T  with reference to  FIG. 1 , and B denotes the number of blocks having the length L, which are used for the differential correlation. 
     If the sync acquirer  620  includes the accumulator  623 , the timing determiner  624  detects the maximum value from among 5×7×N T  values represented by Equation 5, which are stored in the accumulator  623 , and outputs a sample position of the detected maximum value as the detected timing information S 5 . 
       FIG. 9  is a graph illustrating differential correlation values calculated by the differential correlator  621 -A or  621 -B illustrated in  FIG. 8  based on sample positions according to an embodiment of the present invention. For convenience of description, it is assumed that the differential correlation values are obtained in an ideal channel environment in which fading or noise does not exist in a channel between a transmitter end of a base station and the receiver end of a mobile station. 
     In  FIG. 9 , the horizontal axis represents time or a sample index, and the vertical axis represents a differential correlation value at each position of the horizontal axis. Reference numeral  627  denotes a position of a first sample for which the differential correlator  621 -A or  621 -B performs the differential correlation. The differential correlator  621 -A or  621 -B calculates L differential correlation values by obtaining a differential correlation value from the first sample position and provides the calculated L differential correlation values to the accumulator  623 . Thereafter, the differential correlator  621 -A or  621 -B calculates L differential correlation values from a position of a sample next to a sample for which the differential correlator  621 -A or  621 -B performed the last differential correlation and provides the calculated L differential correlation values to the accumulator  623 . The differential correlator  621 -A or  621 -B repeats this process. Among all the L samples, a position at which a peak occurs exists as illustrated in  FIG. 9  as a result of the repetition pattern of sync channel symbols. Here, L denotes a sync block length, i.e. 5×7×N T  with reference to  FIG. 1 . 
       FIG. 10  illustrates a structure of an input signal provided to the group detector  640  illustrated in  FIG. 7  based on sync channel symbol timing acquired by the sync acquirer  620  illustrated in  FIG. 7 , according to an embodiment of the present invention. 
     A CP of each OFDM symbol is removed based on sync channel symbol timing  641  acquired by the sync acquirer  620 , and thereby, N S  sample values are input to the group detector  640  in every sync block. Reference numerals  642 -A,  642 -B,  642 -C,  642 -D, and  642 -E denote positions of sync channel symbols, which are obtained using the acquired sync channel symbol timing  641 . 
       FIG. 11  is a block diagram of the group detector  640  illustrated in  FIG. 7 , according to an embodiment of the present invention. Referring to  FIG. 11 , the group detector  640  includes a frequency offset compensator  645  and a group &amp; boundary detector  650 . 
     The frequency offset compensator  645  sets the sync channel symbol timing  641  based on the output S 5  of the sync acquirer  620 , stores P×N S  received samples ( 642 -A through  642 -E) provided from each of the sync channel band filters  610 -A and  610 -B over several sync block durations based on the sync channel symbol timing  641 , estimates a frequency offset S 8  using the P×N S  received samples ( 642 -A through  642 -E), compensates for frequency offsets of the P×N S  received samples ( 642 -A through  642 -E) based on the estimated frequency offset S 8 , and provides the compensated P×N S  received signal samples S 9  and S 10  to the group &amp; boundary detector  650 . Here, P denotes the number of sync channel symbols used for the frequency offset compensation, the code group detection, and the frame boundary detection, e.g., the number of sync channel symbols included in a single frame, and in this case, P=5 with reference to  FIGS. 1 and 10 . 
     Equations 6 and 7 illustrate frequency offset compensation methods of the frequency offset compensator  645 . 
     
       
         
           
             
               
                 
                   
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     Here, R S  denotes an OFDM sampling frequency, A denotes the number of reception antennas, P denotes the number of sync channel symbols used for the frequency offset compensation, and r a,p (n) denotes an n th  sample value among N S  samples of a p th  sync channel symbol from the initial reference timing  641  provided from the sync acquirer  620  with respect to an a th  reception antenna. 
     Equations 6 and 7 use the repetition pattern of a time domain signal based on the structure of the sync channel symbol in the frequency domain illustrated in  FIG. 3 . In particular, Equation 6 indicates a frequency offset compensation method of a case where a transmitter end transmits a signal without carrying any data on DC subcarriers. 
     Equation 8 illustrates a frequency offset compensation method of the frequency offset compensator  645 . 
     
       
         
           
             
               
                 
                   
                     
                       
                         
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     That is, the frequency offset compensator  645  compensates for frequency offsets of P×N S  received samples as illustrated in  FIG. 10  based on a frequency offset estimated using the frequency offset compensation method illustrated in Equation 8. The frequency offset compensator  645  sequentially provides the frequency offset compensated P×N S  samples S 9  and S 10  (r′ a,p ) to the group &amp; boundary detector  650  in N S  units. 
     The group &amp; boundary detector  650  detects a code group ID and a 10-msec frame timing using the frequency offset compensated samples S 9  and S 10  and the pre-stored hopping code illustrated in Table 1 and provides the detected code group information S 7  and the frame timing information S 6  to the code detector  680  illustrated in  FIG. 7 . 
       FIG. 12  is a block diagram of the group &amp; boundary detector  650  illustrated in  FIG. 11 , according to an embodiment of the present invention. Referring to  FIG. 12 , the group &amp; boundary detector  650  includes code correlation calculators  665 -A and  665 -B, a combiner  656 , a correlation buffer  657 , a hopping code storage unit  659 , a hopping codeword detector  658 , a boundary detector  310 , and a code group detector  300 . 
     Since the mobile station does not know what sync channel sequence index is included in each of the sync channel symbols ( 642 -A through  642 -E), the mobile station must calculate correlations of all possible sequences and N S  samples of each of the sync channel symbols ( 642 -A through  642 -E). 
     The code correlation calculators  665 -A and  665 -B respectively calculate correlations of sync channel sequences and the sync channel symbols S 9  and S 10  frequency offset compensated by the frequency offset compensator  645 . 
     The combiner  656  combines outputs of the code correlation calculators  665 -A and  665 -B and N−1 combined correlation values to the correlation buffer  657  for every sync channel symbol. 
     The correlation buffer  657  buffers N−1 combined correlation values of each P sync channel symbol. That is, P×(N−1) correlation values are stored in the correlation buffer  657 . Here, the minimum value of P is 1. This is because when the hopping codes are orthogonal to the cyclic shifts, a hopping codeword can be detected even when one sync channel sequence is detected. 
     The hopping code storage unit  659  stores a plurality of hopping codewords as illustrated in Table 1. 
     The hopping codeword detector  658  calculates a correlation with each of the stored hopping codewords and all cyclic-shifted codeword of the stored hopping codeword by summing the calculated correlations of the corresponding sync channel sequence and detects a hopping codeword number included in the sync channel symbols, based on the result of the calculation. 
     The boundary detector  310  detects the frame boundary S 6  based on a cyclic shift index of the detected hopping codeword. The boundary detector  310  also detects the code group S 7  based on the detected hopping codeword. A detailed detection process will be described later. 
     In particular, if a sync channel sequence is based on a GCL sequence, the code correlation calculators  665 -A and  665 -B respectively include first data acquirers  800 -A and  800 -B, second data generators  653 -A and  653 -B, and correlation generators  820 -A and  820 -B. 
     The first data acquirers  800 -A and  800 -B respectively acquire data values of positions of subcarriers to which elements of the sync channel sequence are assigned from the frequency offset compensated sync channel symbols S 9  and S 10 . The first data acquirers  800 -A and  800 -B respectively include Fourier transformers  651 -A and  651 -B and demappers  652 -A and  652 -B. The Fourier transformers  651 -A and  651 -B respectively acquire N S  data values by Fourier transforming the sync channel symbols S 9  and S 10 , and each of the demappers  652 -A and  652 -B acquires N data values of subcarriers to which the elements of the sync channel sequence are assigned from among the acquired N S  data values. 
     The second data generators  653 -A and  653 -B respectively receive the outputs of the demappers  652 -A and  652 -B and perform differential encoding defined by Equation 9.
 
 u ( n )= y *( n ) y (( n+ 1) mod N ),  n= 0, 1, . . . ,  N− 1  (9)
 
     Here, y(n) denotes an output of each of the demappers  652 -A and  652 -B, and u(n) denotes an output of each of the second data generators  653 -A and  653 -B. The differential encoding is performed so as to obtain only a linear phase transition corresponding to a GCL sequence number k from N frequency domain signal components. That is, an environment in which channel distortion or noise does not exist is assumed, u(n) is represented by Equation 10. 
     
       
         
           
             
               
                 
                   
                     
                       u 
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     = 
                     
                       exp 
                       ⁢ 
                       
                         { 
                         
                           
                             - 
                             j 
                           
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           2 
                           ⁢ 
                           
                               
                           
                           ⁢ 
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                           ⁢ 
                           
                             n 
                             N 
                           
                           ⁢ 
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                         } 
                       
                     
                   
                   , 
                   
                     
 
                   
                   ⁢ 
                   
                     n 
                     = 
                     0 
                   
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                   ⁢ 
                   
                       
                   
                   , 
                   
                     N 
                     - 
                     1 
                   
                 
               
               
                 
                   ( 
                   10 
                   ) 
                 
               
             
           
         
       
     
     In Equation 10, k denotes a GCL sequence ID, which can have a value from 1 to N−1 as illustrated in Equation 1. 
     The correlation generators  820 -A and  820 -B respectively perform inverse Fourier transformation on N u(n) values of each sync channel symbol, i.e., outputs of the second data generators  653 -A and  653 -B, calculate a correlation of the sync channel symbol and each hopping code with absolute values of the inverse Fourier transformation result. The correlation generators  820 -A and  820 -B respectively include inverse Fourier transformers  654 -A and  654 -B and magnitude calculators  655 -A and  655 -B. 
     The inverse Fourier transformers  654 -A and  654 -B respectively generate N complex samples per sync channel symbol by performing inverse Fourier transformation on the outputs of the second data generators  653 -A and  653 -B. Each of the magnitude calculators  655 -A and  655 -B calculates magnitude of a complex sample by summing a square of a real number component and a square of an imaginary number component for each of the generated N complex samples. In particular, according to an embodiment of the present invention, a first value of the calculated N values is discarded, and only the remaining N−1 values are provided to the combiner  656 . 
       FIG. 13  is a graph illustrating an output of the code correlation calculator  665 -A or  665 -B illustrated in  FIG. 12 , according to an embodiment of the present invention. 
     The horizontal axis represents sync channel sequence (GCL sequence) index, and the vertical axis represents correlation values of currently received sync channel symbols and each of N−1 sync channel sequences (GCL sequences). In particular,  FIG. 13  illustrates an output of the code correlation calculator  665 -A or  665 -B when a hopping codeword element k contained in the currently received sync channel symbol is 2. Referring to  FIG. 13 , a correlation value is largest when k is 2. In particular, if channel distortion or noise does not exist, correlation values at sync channel sequence index positions excluding the sync channel sequence with index k=2 are 0 which is different from the illustration of  FIG. 13 . 
       FIG. 12  is based on the assumption that the mobile station employs reception diversity with two reception antennas, wherein the combiner  656  combines outputs of the code correlation calculators  665 -A and  665 -B, which are acquired according to the reception diversity. If the reception diversity is not used, the combiner  656  and the code correlation calculator  665 -B placed in the lower part can be omitted. 
     The hopping codeword ID respectively corresponds to the code group ID of  FIG. 2 , and the cyclic shift index indicates how far the 10-msec frame boundary is from the first position  641  of the P sync channel symbol durations ( 642 -A through  642 -E) used by the group detector  640  in a unit of sync block length. That is, if the cyclic shift index is 0, the 10-msec frame boundary is the first sync channel symbol position  641  as illustrated in  FIG. 10 , and if the cyclic shift index is 1, the 10-msec frame boundary is a second position  643 -A, and if the cyclic shift index is 2, or 3, the 10-msec frame boundary is a third position  643 -B, or a fourth position  643 -C. 
       FIG. 14  illustrates P×(N−1) correlation values stored in the correlation buffer  657  illustrated in  FIG. 12 , according to an embodiment of the present invention. In  FIG. 14 , it is assumed that P=2 and N=13. The horizontal axis represents sync channel sequence numbers, and the vertical axis represents correlation values of received sync channel symbols and each of N−1 sync channel sequences (GCL sequences). 
     Reference numeral  662 -A indicates correlation values of a first sync channel symbol, i.e., a case of p=0, and 12 sync channel sequences, and each reference numeral  662 -B,  662 -C, or  662 -D indicates correlation values of 12 sync channel sequences calculated for each of sync channel symbols corresponding to p=1, 2, and 3. That is, the uppermost 12 samples  662 -A are an output of the combiner  656  with respect to the first OFDM symbol  642 -A illustrated in  FIG. 10 , the second 12 samples  662 -B are an output of the combiner  656  with respect to the second OFDM symbol  642 -B illustrated in  FIG. 10 , and the third 12 samples  662 -C are an output of the combiner  656  with respect to the third OFDM symbol  642 -C (‘ 642 -A’  ‘ 642 -C’   ) illustrated in  FIG. 10 . The fourth 12 samples  662 -D are also described in the same manner. 
     The hopping codeword detector  658  calculates N G ×P decision variables, selects a decision variable having the maximum value from among the N G ×P decision variables, and provides information on the selected decision variable to the boundary detector  310  and the code group detector  300 . 
     The boundary detector  310  and the code group detector  300  respectively detect the frame boundary S 6  and the code group S 7  based on the provided information. 
     A decision variable w(i) according to an embodiment of the present invention is represented by Equation 11. 
     
       
         
           
             
               
                 
                   
                     
                       w 
                       ⁡ 
                       
                         ( 
                         i 
                         ) 
                       
                     
                     = 
                     
                       
                         ∑ 
                         
                           u 
                           = 
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                           - 
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                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         
                           v 
                           u 
                         
                         ⁡ 
                         
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                             ⁡ 
                             
                               ( 
                               
                                 
                                   ( 
                                   
                                     
                                       i 
                                       
                                         mod 
                                         ⁢ 
                                         
                                             
                                         
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                                     + 
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                                   mod 
                                   ⁢ 
                                   
                                       
                                   
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                               ) 
                             
                           
                           ) 
                         
                       
                     
                   
                   , 
                   
                     
 
                   
                   ⁢ 
                   
                     i 
                     = 
                     0 
                   
                   , 
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                   ⁢ 
                   
                       
                   
                   , 
                   
                     
                       P 
                       × 
                       
                         N 
                         G 
                       
                     
                     - 
                     1 
                   
                 
               
               
                 
                   ( 
                   11 
                   ) 
                 
               
             
           
         
       
     
     Here, mod denotes a modular operator, └x┘denotes the maximum value out of integers equal to or less than x, N G  denotes the number of code groups or hopping codewords used in a system, which is indicated to be 3 based on Table 1, P denotes a hopping codeword length or the number of sync channel symbols per 10-msec frame, which is indicated to be 4 based on  FIG. 1  and Table 1, and h x (y) denotes an index value of a y th  element of a hopping codeword whose index is x, e.g., when x=0 and y=2, h 0(2)  is 9 with reference to Table 1. 
     In Equation 11, v u (k) is a correlation value of a sync channel sequence whose index is k with respect to a u th  OFDM symbol position and is stored in the correlation buffer  657 . Equation 11 represents a decision variable of each of the hopping codewords of Table 1 and all cyclically shifted codewords of the hopping codewords. That is, a decision variable of a hopping codeword {3, 6, 9, 12} whose index is 0 is w(0), a decision variable of a hopping codeword {9, 5, 6, 7, 8}, which is “1” cyclically shifted from the hopping codeword {6, 9, 12, 3} whose index is 0, is w(1), and a decision variable of a hopping codeword, which is “u” cyclically shifted from a hopping codeword whose index is i, is w(i×P+u). 
     A process of calculating w(i) by referring to  FIG. 14  and Table 1 will now be described in detail. w(0) is a decision variable of a hopping codeword {3, 6, 9, 12} whose ID is 0 and cyclic shift index is 0, i.e., w(0)=2.2+1.9+1.5+0.6=5.1. w(2) is a decision variable of a hopping codeword {9, 12, 3, 6} whose ID is 0 and cyclic shift index is 2, i.e., w(2)=2.2+1.6+1.7+30.1=8.6. Through this process, w(0), w(1), through to w(P×N G −1) are calculated, and if w(10=2×4+2) has the maximum value with 10.2=9.6=8.0=9.1=36.9, the hopping codeword detector  658  finally determines that a hopping code index is 2 and a cyclic shift index is 2, and detects a frame boundary and a code group. 
     That is, if it is assumed that an index of a decision variable having the maximum value among P×N G −1 decision variables, i.e., w(0), w(1), through to w(P×N G −1), is i max , i.e., 
                 i   max     =       max   i     ⁢     w   ⁡     (   i   )           ,         
the hopping codeword detector  658  calculates a hopping code index and a cyclic shift index as └i max ÷P┘ and (i max ) mod P′  respectively. Since a hopping codeword respectively corresponds to a code group, a code group can be detected from an index of a hopping codeword, and a frame boundary can be detected from a cyclic shift index.
 
     According to an embodiment of the present invention, decision variable information provided from the hopping codeword detector  658  to the boundary detector  310  and the code group detector  300  is i max . The boundary detector  310  detects a cyclic shift index by performing a modular operation (i max ) mod P  using the received decision variable information i max  and detects a frame boundary based on the detected cyclic shift index. The code group detector  300  detects an index of a hopping codeword by performing an operation └i max ÷P┘ using the received decision variable information i max  and detects a code group corresponding to the detected hopping code index. 
     Meanwhile, although in regard to the current embodiment a description is given of the processes of detecting the frame boundary and the code group based on when P=4, that is, based on the number of sync channel symbols in one frame, the same processes can be applied when P=1, using the orthogonality of the hopping codes and the cyclic shifts. A process of searching a cell using only the first sync channel symbol based on a sync channel symbol sync obtained in the first step of cell search process will now be described. First, an output of the combiner  656  is 12 correlation values of the first sync channel symbol, and when these correlation values correspond to each value of 662-A as shown in  FIG. 14 , the peak correlation value from among the 12 correlation values occurs on the position of the sync channel sequence index value 8. That is, a hopping codeword element  8  is detected, and it can be presumed that the hopping codeword element  8  belongs to hopping codeword  2  and that the cyclic shift index is 2, using Table 1. 
     As described above, if each code group contains only one scrambling code, a hopping codeword respectively corresponds to a scrambling code, and thus the hopping codeword detector  658  can detect a scrambling code from the detected code group. Thus, in this case, the third step of the cell search process can be omitted or used for verification of a scrambling code detected in the second step of the cell search process. 
       FIG. 15  illustrates a structure of frame information acquired in the second step of the cell search process according to an embodiment of the present invention. The code detector  680  detects a scrambling code based on the frame information. 
     Referring to  FIG. 15 , reference numeral  675  denotes a frame boundary detected in the second step of the cell search process, i.e., by the group detector  640 , and the code detector  680  can acquire positions of common pilot channel symbols, i.e., common pilot channel symbol durations  678 , based on the detected frame boundary  675  and can finally detect a scrambling code of a target base station by performing pilot correlation between the common pilot channel symbol and scrambling codes belonging to a code group detected in the second step of the cell search process based on the acquired position. Each common pilot channel symbol includes N T  samples as other OFDM samples, including a CP duration having N CP  samples and a remainder duration  679  having N S  samples. 
     In other words, the code detector  680  extracts a common pilot channel symbol contained in a received sub-frame based on the frame boundary information acquired in the second step of the cell search process, calculates correlation values of the extracted common pilot channel symbol and scrambling codes belonging to a code group detected in the second step of the cell search process, and determines a scrambling code of a current base station with a scrambling code corresponding to the maximum correlation value out of the calculated correlation values. That is, the common pilot channel symbol is used to estimate a channel for coherent demodulation of a data channel of a forward link and detect a scrambling code in the third step of the cell search process according to an embodiment of the present invention. 
     Complexity of the receiver can be reduced by the code detector  680  searching only scrambling codes belonging to a code group received from the group detector  640 . That is, as illustrated in  FIG. 2 , at least 19 scrambling codes exist in the system, and the code detector  680  can search only N c  scrambling codes belonging to a code group detected in the second step of the cell search process among the 512 scrambling codes. Here, N c  denotes the number of scrambling codes per code group, N c =4 according to  FIG. 2 . 
       FIG. 16  is a block diagram of the code detector  680  illustrated in  FIG. 7 , according to an embodiment of the present invention. Referring to  FIG. 16 , the code detector  680  includes frequency offset compensators  681 -A and  681 -B, Fourier transformers  682 -A and  682 -B, pilot symbol extractors  683 -A and  683 -B, pilot correlators  684 -A and  684 -B, accumulators  686 -A and  686 -B, a combiner  687 , and a peak detector  688 . 
     Since each of the frequency offset compensators  681 -A and  681 -B can detect the common pilot channel symbol duration  678  of each sub-frame based on the 10-msec frame boundary information S 6   675  received from the group detector  640 , each of the frequency offset compensators  681 -A and  681 -B frequency offset compensates the N S  samples  679  excluding the CP of the common pilot channel symbol contained in the down-converted OFDM symbols S 1  or S 2  using Equation 8. Here, the frequency offset estimation value S 8  received from the group detector  640  can be used as a frequency offset estimation value used for the frequency offset compensation. 
     Each of the Fourier transformers  682 -A and  682 -B generates a frequency domain signal by performing Fourier transformation on the N S  frequency offset compensated samples. 
     Each of the pilot symbol extractors  683 -A and  683 -B extracts only N, pieces of pilot symbol from the generated frequency domain signal. 
     Each of the pilot correlators  684 -A and  684 -B calculates correlation values of the extracted N P  pieces of pilot symbol and the N c  scrambling codes belonging to the code group received from the group detector  640 . Here, Equations 12 through 15 can be used to calculate the correlation values. Each of the pilot correlators  684 -A and  684 -B includes N c  differential correlators performing a differential correlation operation in a parallel method. That is, each of the N c  differential correlators calculates a correlation value of each extracted pilot symbol and each scrambling code belonging to the code group. Each of the N c  differential correlators operates in the common pilot channel symbol duration  678  of each sub-frame, and an output of each of the N c  differential correlators is accumulated in each sub-frame accumulator included in the accumulators  686 -A and  686 -B based on the N c  scrambling codes belonging to the detected code group. Equation 12 through 15 will be described later. 
     Each of the accumulators  686 -A and  686 -B accumulates N c  correlation values calculated with respect to each common pilot channel symbol. Referring to  FIG. 1 , since one common pilot channel symbol per sub-frame exists, each of the accumulators  686 -A and  686 -B accumulates correlation values corresponding to a predetermined number of sub-frames. Each of the accumulators  686 -A and  686 -B includes N c  sub-frame accumulators. 
     The combiner  687  generates N c  decision variables by combining outputs of the accumulators  686 -A and  686 -B, which are calculated through a plurality of paths according to the reception diversity realized using a plurality of antennas. It will be understood by those of ordinary skill in the art that the combiner  687  and the blocks in the lower part can be omitted if reception diversity is not used. 
     The peak detector  688  finally detects a scrambling code S 11  of a current base station by detecting a decision variable having the maximum value out of the N c  decision variables provided by the combiner  687  and selecting a scrambling code corresponding to the detected decision variable. Through this process, the mobile station can detect a scrambling code of a base station having the shortest radio distance or a base station having the highest reception signal intensity. If the detected maximum value is greater than a predetermined threshold, it is considered that the cell search process has been completed, and if the detected maximum value is less than the predetermined threshold, the cell search unit  600  according to an embodiment of the present invention repeatedly performs the first, second, and third steps of the cell search process. 
     If each code group contains only one scrambling code, i.e., if N c =1, a code group ID respectively corresponds to a scrambling code ID, and thus a frame boundary and a scrambling code ID, which are the purpose of the present invention, can be detected even when only the first and second steps of the cell search process are performed. Thus, in this case, the third step of the cell search process can be omitted or used for verification of a scrambling code ID detected in the second step of the cell search process. 
     An operation of the pilot correlator  684 -A or  684 -B will now be described in detail. 
       FIG. 17  is a conceptual diagram for explaining an operation of the pilot correlator  684 -A or  684 -B according to an embodiment of the present invention. 
     Referring to  FIG. 17 , reference numerals  695  and  696  respectively denote an input and an output of the pilot symbol extractors  683 -A or  683 -B. That is, pilot symbol and data symbol may exist together in a frequency domain signal, and in this case, the pilot symbol extractor  683 -A or  683 -B extracts N P  pieces of pilot symbol from the frequency domain signal  695 . In  FIG. 17 , X(n) denotes n th  pilot symbol from among frequency domain data of a common pilot channel symbol. In particular, in  FIG. 17 , the common pilot channel symbol contains N P  pieces of pilot symbol. 
     Equations 12 through 15 represent a method of correlating the extracted pilot symbol and a scrambling code. 
     
       
         
           
             
               
                 
                   
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                                         + 
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                     } 
                   
                 
               
               
                 
                   ( 
                   12 
                   ) 
                 
               
             
           
         
       
     
     Here, N P  denotes the number of pieces of pilot symbol in the frequency domain, which are contained in a common pilot channel symbol, and c g     k   (u) denotes a u th  element of a k th  scrambling code out of scrambling codes belonging to a detected code group. 
     The differential correlation represented by Equation 12 is used in the third step of the cell search process according to an embodiment of the present invention because of the following reason. In an OFDM signaling method, adjacent symbols in the frequency domain undergo almost the same wireless fading. That is, channel distortion over the adjacent symbols is almost the same. However, wireless fading between symbols far from each other is independent if a gap between the symbols in the frequency domain is large. In this case, the performance of a conventional frequency domain correlator defined by Equation 13 is significantly decreased if a correlation length N is large. 
     
       
         
           
             
               
                 
                   
                     ∑ 
                     
                       i 
                       = 
                       0 
                     
                     
                       N 
                       - 
                       1 
                     
                   
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   
                     { 
                     
                       ( 
                       
                         
                           X 
                           ⁡ 
                           
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                             i 
                             ) 
                           
                         
                         ⁢ 
                         
                           
                             ( 
                             
                               c 
                               ⁡ 
                               
                                 ( 
                                 i 
                                 ) 
                               
                             
                             ) 
                           
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                       ) 
                     
                     } 
                   
                 
               
               
                 
                   ( 
                   13 
                   ) 
                 
               
             
           
         
       
     
     That is, since X(i)=a i c(i) in Equation 13, Equation 13 becomes 
                 ∑     i   =   0       N   -   1       ⁢           ⁢     a   i       ,         
and thus, a wireless fading effect is coherently added for independent symbols X( ) far from each other, and the performance of the conventional frequency domain correlator is significantly decreased in a channel that undergoes fading. Here, a i  denotes a channel value of an i th  subcarrier and has a characteristic in that channel values are almost the same for adjacent subcarriers in the fading channel but different from each other for subcarriers far from each other.
 
     
       
         
           
             
               
                 
                   
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                           ( 
                           
                             
                               X 
                               ⁡ 
                               
                                 ( 
                                 
                                   
                                     2 
                                     ⁢ 
                                     i 
                                   
                                   + 
                                   1 
                                 
                                 ) 
                               
                             
                             ⁢ 
                             
                               
                                 ( 
                                 
                                   c 
                                   ⁡ 
                                   
                                     ( 
                                     
                                       
                                         2 
                                         ⁢ 
                                         i 
                                       
                                       + 
                                       1 
                                     
                                     ) 
                                   
                                 
                                 ) 
                               
                               * 
                             
                           
                           ) 
                         
                         * 
                       
                     
                     } 
                   
                 
               
               
                 
                   ( 
                   14 
                   ) 
                 
               
             
           
         
       
     
     However, if a differential correlator defined by Equation 14 is used, a correlation value becomes 
                   ∑     i   =   0         N   2     -   1       ⁢       a     2   ⁢   i       ⁢     a       2   ⁢   i     +   1     *         ≈       ∑     i   =   0         N   2     -   1       ⁢            a     2   ⁢   i            2         ,         
and thus a better performance can be achieved than the conventional frequency domain correlator.
 
     In the third step of the cell search process according to an embodiment of the present invention, the reason differential multiplication is used between every other pilot symbol as in Equation 12 or reference numeral  697  as illustrated in  FIG. 17  instead of using differential multiplication between adjacent symbols as in Equation 14 is due to the fact that the mobile station cannot detect information about a current base station to which the mobile station belongs in an initial sync acquisition mode. That is, the mobile station cannot detect whether the number of transmission antennas used in the current base station is 1 or 2. 
     If the number of transmission antennas is 1, all pilot symbols  696  illustrated in  FIG. 17  are transmitted through the same transmission antenna. However, if the number of transmission antennas is 2, even-th pilot symbols (i.e., X(0), X(2), . . . ) are transmitted through a first transmission antenna, and odd-th pilot symbols are transmitted through a second transmission antenna. In this case, i.e., if the number of transmission antennas is 2, pilot symbols that are adjacent in the frequency domain undergo fully independent fading. In this case, if a receiver end performs differential multiplication between adjacent pilot symbols as in Equation 14, detection performance may be decreased. However, if the differential correlation according to an embodiment of the present invention is performed as illustrated by reference numeral  697  of  FIG. 17 , i.e., if differential multiplication  697 -A between even-th pilot symbols and differential multiplication  697 -B between odd-th pilot symbols are performed, a long PN scrambling code ID can be detected regardless of whether the number of transmission antennas used in the current base station is 1 or 2. In order to reduce complexity, Equation 12 can be replaced by Equation 15 by using only the even-th pilot symbol and ignoring the odd-th pilot symbol. 
     
       
         
           
             
               
                 
                   
                     ∑ 
                     
                       i 
                       = 
                       0 
                     
                     
                       
                         
                           N 
                           p 
                         
                         4 
                       
                       - 
                       1 
                     
                   
                   ⁢ 
                   
                     { 
                     
                       
                         ( 
                         
                           
                             X 
                             ⁡ 
                             
                               ( 
                               
                                 4 
                                 ⁢ 
                                 i 
                               
                               ) 
                             
                           
                           ⁢ 
                           
                             
                               ( 
                               
                                 
                                   c 
                                   
                                     g 
                                     k 
                                   
                                 
                                 ⁡ 
                                 
                                   ( 
                                   
                                     4 
                                     ⁢ 
                                     i 
                                   
                                   ) 
                                 
                               
                               ) 
                             
                             * 
                           
                         
                         ) 
                       
                       ⁢ 
                       
                         
                           ( 
                           
                             
                               X 
                               ⁡ 
                               
                                 ( 
                                 
                                   
                                     4 
                                     ⁢ 
                                     i 
                                   
                                   + 
                                   2 
                                 
                                 ) 
                               
                             
                             ⁢ 
                             
                               
                                 ( 
                                 
                                   
                                     c 
                                     
                                       g 
                                       k 
                                     
                                   
                                   ⁡ 
                                   
                                     ( 
                                     
                                       
                                         4 
                                         ⁢ 
                                         i 
                                       
                                       + 
                                       2 
                                     
                                     ) 
                                   
                                 
                                 ) 
                               
                               * 
                             
                           
                           ) 
                         
                         * 
                       
                     
                     } 
                   
                 
               
               
                 
                   ( 
                   15 
                   ) 
                 
               
             
           
         
       
     
     When the mobile station is turned on, an error of the clock generator  540  may be 3 pulses per million (PPM) or more. If this error is converted to a value used in a 2 GHz band, the error is 6 KHz or more. If a frequency offset is large in the initial cell search process, the search performance in the first step of the cell search process may be significantly decreased. There is no performance problem in the second and third steps of the cell search process since frequency offset compensation is performed. 
       FIG. 18  is a block diagram of the sync acquirer  620  illustrated in  FIG. 7 , according to another embodiment of the present invention. Referring to  FIG. 18 , the sync acquirer  620  includes frequency offset switching units  700 -A and  700 -B, differential correlators  721 -A and  721 -B, an accumulator  723 , and a timing determiner  724 . Since functions and operations of the differential correlators  721 -A and  721 -B, the accumulator  723 , and the timing determiner  724  are the same as those illustrated in  FIG. 8 , a detailed description thereof is omitted, and only the frequency offset switching units  700 -A and  700 -B will be described. 
     If a correlation operation handling absolute values is performed as in Equation 3 or 4, no frequency offset effect can be considered. However, if a correlation operation different from Equation 3 or 4 is performed, the frequency offset switching units  700 -A and  700 -B according to an embodiment of the present invention may be further included. 
     The frequency offset switching unit  700 -A or  700 -B multiplies an input signal r(n) by an arbitrary frequency offset component as in Equation 16, wherein a different offset value is used in every unit duration during the first step of the cell search process. 
     
       
         
           
             
               
                 
                   
                     
                       
                         r 
                         ′ 
                       
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     = 
                     
                       
                         r 
                         ⁡ 
                         
                           ( 
                           n 
                           ) 
                         
                       
                       × 
                       exp 
                       ⁢ 
                       
                         { 
                         
                           
                             - 
                             j2 
                           
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           π 
                           ⁢ 
                           
                             
                               Δ 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 f 
                                 s 
                               
                             
                             
                               R 
                               s 
                             
                           
                           ⁢ 
                           n 
                         
                         } 
                       
                     
                   
                   , 
                   
                     n 
                     = 
                     0 
                   
                   , 
                   1 
                   , 
                   2 
                   , 
                   … 
                 
               
               
                 
                   ( 
                   16 
                   ) 
                 
               
             
           
         
       
     
       FIG. 19  is a conceptual diagram for explaining an operation of the frequency offset switching unit  700 -A or  700 -B illustrated in  FIG. 18  according to an embodiment of the present invention. 
     The upper part of  FIG. 19  illustrates frequency offset values 0 KHz, −6 KHz, and 6 KHz used in the frequency offset switching unit  700 -A or  700 -B. In this case, the sync acquirer  620  can safely operate even with an initial frequency offset of more than 18 KHz. 
     The lower part of  FIG. 19  illustrates five cell search durations, wherein a frequency offset component used in the first step of the cell search process is illustrated in each cell search duration. Each cell search duration is 10 msec as shown in  FIG. 19 . 
       FIG. 20  is a flowchart illustrating a cell search method of a mobile station according to an embodiment of the present invention. 
     Referring to  FIG. 20 , the mobile station&#39;s cell search method according to the current embodiment includes operations sequentially processed by the cell search unit  600  illustrated in  FIG. 7 . Thus, although not fully described, the contents relating to the cell search unit  600  also apply to the cell search method according to the current embodiment. 
     The sync acquirer  620  acquires sync block synchronization using a sync channel symbol of a forward link in operation S 200 . 
     In operation S 210 , the group detector  640  detects a hopping codeword and a cyclic shift contained in a forward link frame based on the acquired synchronization and detects a code group to which a scrambling code of a current base station belongs and a frame boundary based on the detected hopping codeword and cyclic shift. 
     In operation S 220 , the code detector  680  detects the scrambling code based on the detected code group and a common pilot channel symbol. 
     In operation S 230 , it is determined whether the detected scrambling code is reliable, and if it is determined that the detected scrambling code is not reliable, the process returns to operation S 200  and performs sync acquisition of a subsequent observing duration. If it is determined that the detected scrambling code is reliable, the detected scrambling code is considered to be a scrambling code of the home cell and the cell search process ends. A fine tuning operation for fine tuning frequency and timing can be further included after operation S 230 . A method of determining whether a correlation value used to detect the scrambling code is less than a predetermined threshold value can be used as a method of determining reliability. 
     If each code group includes only one scrambling code, the scrambling code can be detected in operation S 210  since a hopping codeword respectively corresponds to a scrambling code. Thus, in this case, operation S 220  can be omitted. If operation S 220  is performed, the scrambling code detected in operation S 220  is used for verification of the scrambling code detected in operation S 210 . That is, in this case, the scrambling code detected in operation S 220  becomes a determination reference for determining reliability in operation S 230 . According to this method, if the two detected scrambling codes are different from each other, the process can return to operation S 200  after operation S 220 . 
       FIG. 21  is a flowchart illustrating a forward link frame transmission method of a base station according to an embodiment of the present invention. Referring to  FIG. 21 , the base station&#39;s forward link frame transmission method according to the current embodiment includes operations sequentially processed by the blocks illustrated in  FIG. 5 . Thus, although not fully described, the contents described relating to the base station illustrated in  FIG. 5  also apply to the forward link frame transmission method according to the current embodiment. 
     In operation S 400 , the sync channel generator  400  generates a hopping codeword specifying a code group to which a scrambling code of the base station belongs, and the OFDM symbol mappers  404 -A and  404 -B perform sync channel sequence hopping by assigning each element of the generated hopping codeword to each sync channel symbol. Simultaneously, data generated by the data channel generator  402  and the common pilot channel generator  401  is assigned to each position of the frequency domain by the OFDM symbol mappers  404 -A and  404 -B. 
     In operation S 410 , symbols that remain due to the exclusion of the sync channel symbols are scrambled in the frequency domain by the scramblers  405 -A and  405 -B. 
     In operation S 420 , a forward link frame is generated by performing inverse Fourier transformation on each sync channel symbol and scrambled remaining symbol in the inverse Fourier transformers  406 -A and  406 -B and inserting CPs into the forward link frame in the CP insertion units  407 -A and  407 -B. 
     In operation S 430 , the generated forward link frame is transmitted through an RF channel by the IF/RF units  408 -A and  408 -B. 
     As described above, according to the present invention, in an OFDM cellular system, a cell search time of a mobile station can be reduced, and a cell search unit operating with low complexity can be implemented. In addition, with only one sync channel, OFDM symbol synchronization, a long scrambling code group ID, and a frame boundary can be detected, and frequency offset estimation can be performed. 
     In addition, according to a sync acquisition method, synchronization can be acquired with low complexity. 
     In addition, in an OFDM cellular system in which base stations are in a sync mode, an adjacent cell search process can be efficiently performed, and thus handover can be smoothly performed, and battery consumption of a mobile station can be reduced. 
     The invention can also be embodied as computer readable codes on a computer readable recording medium. The computer readable recording medium is any data storage device that can store data which can be thereafter read by a computer system. Examples of the computer readable recording medium include read-only memory (ROM), random-access memory (RAM), CD-ROMs, magnetic tapes, floppy disks, optical data storage devices, and carrier waves (such as data transmission through the Internet). The computer readable recording medium can also be distributed over network coupled computer systems so that the computer readable code is stored and executed in a distributed fashion. Also, functional programs, codes, and code segments for accomplishing the present invention can be easily construed by programmers skilled in the art to which the present invention pertains. 
     While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it will be understood by those of ordinary skill in the art that various changes in form and details may be made therein without departing from the spirit and scope of the present invention as defined by the following claims.