Patent Publication Number: US-6658354-B2

Title: Interruption free navigator

Description:
BACKGROUND OF THE PRESENT INVENTION 
     1. Field of the Present Invention 
     The present invention relates to a positioning method and system, and more particularly to an interruption flee navigator for both stationary and hand-held applications, which can ensure the user be provided with accurate positioning data continuously anytime and anywhere, no matter the user is located inside a building, tunnel, forested area, unbanized terrain, or high jamming environment where GPS signals are normally not available. 
     2. Description of Related Arts 
     Current hand-held navigator depend on the GPS (Global Positioning System). The GPS is a satellite radio-navigation system, which is owned, deployed, and operated by the U.S. Department of Defense, but is available for commercial uses around the world. Unfortunately, GPS is vulnerable to jamming and shadowing, especially in hand-held application on hand, so that a GPS receiver often can not provide continuous positioning information without interruption, especially in urban area. For commercial hand-held applications, it is very important to obtain continuous positioning information in urban area. On-the-fly ambiguity resolution is one the most recent addition to the GPS to toolbox to determine the correct number of initial integer cycles in carrier-phase measurements, while a receiver is in motion, as indicated, in another patent application of the applicant application Ser. No. 09/611,568. 
     Traditionally, an inertial navigation system (INS) is an interruption-free navigation system, which does not need to receive any external radio frequency signals to Output position data continuously. However, the cost, size, power, and position-drift characteristics of conventional inertial navigation systems prohibit them from use in commercial hand-held navigation applications. 
     Therefore, it is very desirable to develop a navigator with reasonable size and weight and power consumption for hand-held operation without interruption, which can be used in the areas where GPS signals are not available, such as inside buildings, tunnels, forested areas, urbanized terrain, and high jamming environments. 
     Moreover, it is well known that the position errors of an inertial navigation system drift with time and GPS has a poor accuracy of vertical positioning. The long term accuracy of GPS and inertial navigation system integration solution mainly depends on the performance of GPS. It means that the GPS and inertial integrated navigation system can not improve the vertical positioning performance. For some applications, such as commercial and military operations inside a skyscraper, altitude accuracy and environment situation data are very critical. 
     SUMMARY OF THE PRESENT INVENTION 
     The main objective of the present invention is to provide an interruption free navigator to determine position information of a user with high accuracy, wherein the present invention can ensure the user be provided with accurate positioning data continuously anytime and anywhere, no matter the user is located inside a building, tunnel, forested area, urbanized terrain, or high jamming environment where GPS signals are normally not available. In order to accomplish the main objective, the interruption free navigator comprises a main IMU based interruption-free positioning module for sensing motion measurements of the user and produce interruption-free positioning data of the user, a positioning assistant which is incorporated to provide interrupted or continuous positioning data to assist the main IMU based interruption-free positioning module to achieve an improved interruption-free positioning data of the user, a wireless communication device which is built in to exchange the user position information with other users, and a display device which is attached to provide location and surrounding information by selectively accessing a map database with the user position information 
     Another objective of the interruption free navigator of the present invention is to determine position information of a user with high accuracy. The system of the present invention can receive but not rely on GPS signals and DGPS (differential GPS) signals for a highly accurate positioning solution. Without GPS/DGPS signals, the system also provides a highly accurate positioning solution, such as an accuracy of better than 1 percent of the distance traveled. The system is a right positioning system with reasonable size and weight and power consumption for commercial stationary or hand-held operation, which can be used in areas where GPS signals are not available, such as tunnels, forested areas, urbanized terrain, and high jamming environments. 
     Another objective of the interruption free navigator of the present invention is to determine position information of a user with high accuracy, wherein an altitude measurement device and/or an object detection system are incorporated to further meet other needs such as improving the vertical positioning performance. 
     Another objective of the interruption free navigator of the present invention is to determine position information of a user on land with high accuracy, such as an accuracy of better than 1 percent of the distance traveled without relying on GPS, wherein output signals of an inertial measurement unit, a velocity producer, and a north finder are processed to obtain highly accurate position measurements of the user on land. 
     Another objective of the interruption free navigator of the present invention is to determine position information of a user with high accuracy, wherein a wireless communication device is built in to exchange the user position information with other users. 
     Another objective of the interruption free navigator of the present invention is to determine position information of a user, wherein a display device is employed to provide location and surrounding information by selectively accessing a map database with the user position information. 
     Another objective of the interruption free navigator of the present invention is to fuse information from the IMU, a velocity producer, and a north finder to achieve a highly accurate interruption-free navigation solution with hardware and software modules, including the following capabilities: 
     (a) Interruption-free navigation. 
     (b) Autonomous position error ≦1% of the distance traveled when GPS RF (radio frequency) signals are not available. 
     (c) Low cost, low power consumption, lightweight. 
     (d) A unique sophisticated Kalman filter. It removes the inherent drift of the free inertial positioning solution derived from the output of the low cost “core micro” IMU by means of fusing information from the “core micro” IMU, magnetic heading sensor, and velocity producer. 
     (e) Smoothing of the output noise of the magnetic sensor and velocity producer. 
     (f) Innovative state variable selection and measurement design of the Kalman filter, 
     including position update, relative position update, heading update, and automated zero velocity update. 
     (g) Autonomous multiple user stop tests and associated zero-velocity updates. The user stop is not required to perform a zero-velocity update. But, if the user stops at will, the system can exploit such a benefit autonomously. 
     (h) Advanced IMU—MEMS (MicroElectroMechanicalSystems) and ASIC (Application Specific Integrated Circuit) based “core micro” IMU: Miniaturized (Length/Width/Height) and Lightweight; High Performance and Low Cost; Low Power Dissipation; Exceptional Improvement in Reliability. 
     (i) Map database and software module to access surrounding information using the current position solution. 
     (j) Display device and software module to visualize the location of the user and the surrounding information. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram illustrating the system configuration according to a preferred embodiment of the present invention. 
     FIG. 2A is a block diagram illustrating a first alternative mode of the above preferred embodiment of the system configuration of the present invention. 
     FIG. 2B is a block diagram illustrating a second alternative mode of the above preferred embodiment of the system configuration of the present invention. 
     FIG. 3 shows characteristics comparison of a pure INS and an aided INS. 
     FIG. 4 shows characteristics of temperature induced error of a MEMS angular rate sensor. 
     FIG. 5 is a block diagram illustrating the positioning assistant of the system configuration is a GPS receiver according to the above preferred embodiment of the present invention. 
     FIG. 6 is a block diagram illustrating the processing modules of the navigation processor of the system configuration of the above preferred embodiment of the present invention. 
     FIG. 7 is a block diagram illustrating the positioning assistant of the system configuration including both the GPS receiver and a data link according to the above preferred embodiment of the present invention. 
     FIG. 8 is a block diagram illustrating the processing modules of the navigation processor including DGPS according to the above preferred embodiment of the present invention. 
     FIG. 9 is a block diagram illustrating processing modules of the inertial navigation processing according to the above preferred embodiment of the present invention. 
     FIG. 10 is a block diagram illustrating the processing of the flux valve according to the above preferred embodiment of the present invention. 
     FIG. 11 is a block diagram illustrating the Doppler radar processing according to the above preferred embodiment of the present invention. 
     FIG. 12 is a block diagram illustrating the computation of the relative position measurements according to the above preferred embodiment of the present invention. 
     FIG. 13 is a block diagram illustrating the computation of the Kalman filter according to the above preferred embodiment of the present invention. 
     FIG. 14 is a flow diagram of the new process for on-the-fly ambiguity resolution technique of the present invention. 
     FIG. 15 is a flow diagram of intermediate ambiguity search strategy (IASS) according to the new process for the on-the-fly ambiguity resolution technique of the present invention. 
     FIG. 16 is a block diagram of the procedure for forming the estimator bank according to the new process for no-the-fly ambiguity resolution technique of the present invention. 
     FIG. 17 is a complete form of the estimator bank according to the new process for the on-the-fly ambiguity resolution technique of the present invention. 
     FIG. 18 is a block diagram illustrating the processing module for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 19 is a block diagram illustrating the processing modules with thermal control processing for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 20 is a block diagram illustrating the processing modules with thermal compensation processing for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 21 is a block diagram illustrating an angular increment and velocity increment producer for outputting voltage signals of the angular rate producer and acceleration producer for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 22 is a block diagram illustrating another angular increment and velocity increment producer for outputting voltage signals of angular rate producer and acceleration producer for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 23 is a block diagram illustrating another angular increment and velocity increment producer for outputting voltage signals of an angular rate producer and acceleration producer for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 24 is a block diagram illustrating another angular increment and velocity increment producer for outputting voltage signals of an angular rate producer and acceleration producer for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 25 is a block diagram illustrating a thermal processor for outputting analog voltage signals of the thermal sensing producer according to the above preferred embodiment of the present invention. 
     FIG. 26 is a block diagram illustrating another thermal processor for outputting analog voltage signals of the thermal sensing producer according to the above preferred embodiment of the present invention. 
     FIG. 27 is a block diagram illustrating another thermal processor for outputting analog voltage signals of the thermal sensing producer according to the above preferred embodiment of the present invention. 
     FIG. 28 is a block diagram illustrating a processing module for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 29 is a block diagram illustrating a temperature digitizer for outputting analog voltage signals of the thermal sensing producer according to the above preferred embodiment of the present invention. 
     FIG. 30 is a block diagram illustrating a temperature digitizer for outputting analog voltage signals of the thermal sensing producer according to the above preferred embodiment of the present invention. 
     FIG. 31 is a block diagram illustrating a processing module with thermal compensation processing for the coremicro inertial measurement unit according to the above preferred embodiment of the present invention. 
     FIG. 32 is a block diagram illustrating the attitude and heading processing module according to the above preferred embodiment of the present invention. 
     FIG. 33 is a functional block diagram illustrating the position velocity attitude and heading module according to the above preferred embodiment of the present invention. 
     FIG. 34 is a perspective view illustrating the inside mechanical structure and circuit board deployment in the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 35 is a sectional side view of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 36 is a block diagram illustrating the connection among the four circuit boards inside the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 37 is a block diagram of the front-end circuit in each of the first, second, and fourth circuit boards of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 38 is a block diagram of the ASIC chip in the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 39 is a block diagram of processing modules running in the DSP chipset in the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 40 is a block diagram of the angle signal loop circuitry of the ASIC chip in the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 41 is block diagram of the dither motion control circuitry of the ASIC chip in the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 42 is a block diagram of the thermal control circuit of the ASIC chip in the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
     FIG. 43 is a block diagram of the dither motion processing module running in the DSP chipset of the third circuit board of the coremicro IMU according to the above preferred embodiment of the present invention. 
    
    
     DETAIL DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Referring to FIG. 1, an interruption free navigator of the present invention comprises a main IMU based interruption-free positioning module  10  a positioning assistant  8 , a wireless communication device  4 , a map database  5 , and a display device. 
     The main IMU based interruption-free positioning module  10  is utilized for sensing motion measurements of the user by the IMU and producing interruption-free positioning data of the user. 
     The positioning assistant  8  is adapted for providing interruptible positioning data to assist the main IMU based interruption-free positioning module to achieve an improved interruption-free positioning data of the user. 
     The wireless communication device  4  is built in for exchanging the improved interruption-free positioning data with other users. 
     The map database  5  is selectively adapted for providing map data to obtain surrounding map information of location of the user by accessing the map database using the improved interruption-free positioning data. 
     The display device  7  is for visualizing the improved interruption-free positioning data of the user using the surrounding map information. 
     Referring to FIG. 5, the main IMU based interruption-free positioning module  10  further comprises an inertial measurement unit (IMU)  1 , a north finder  2 , a navigation processor  3 , and a velocity producer  6 .: 
     The inertial measurement unit (IMU) is equipped for sensing the traveling displacement motions of the user so as to produce digital angular increments and velocity increments data in response to the user motion. The north finder  2  produces the heading measurement of the user. The velocity producer  6  produces velocity data in the body frame of the user. 
     The navigation processor  3  is connected with the inertial measurement unit  1 , the north finder  2 , the velocity producer  6 , and the positioning assistant  8 , so as to receive the digital angular increments and velocity increments data, the heading measurement, the velocity data in the body frame, and the interruptible positioning data from the positioning assistant  8  to produce IMU position, velocity, and attitude data, and an optimal estimate of errors of IMU position, velocity, and attitude data for correcting the IMU position, velocity, and attitude data error to output the corrected IMU position, velocity and attitude data by means of a real-time software. 
     The real-time software comprises an inertial navigation processing module for producing the IMU position, velocity, and attitude data, and an optimal filtering module for producing the optimal estimate of errors of IMU position, velocity, and attitude data; wherein the optimal estimate of errors of IMU position, velocity, and attitude are used to correct the IMU position, velocity, and attitude data error to output the corrected IMU position, velocity, and attitude data. 
     It is worth to mention that rapid advance in MEMS technologies makes it possible to fabricate low cost, light weight, miniaturized size, and low power angular rate sensors and accelerometers, “MEMS” stands for “MicroElectroMechanical Systems”, or small integrated electrical/mechanical devices. MEMS devices involve creating controllable mechanical and movable structures using IC (Integrated Circuit) technologies. MEMS includes the concepts of integration of Microelectronics and Micromachining. Examples of successful MEMS devices include inkjet-printer cartridges, accelerometers that deploy car airbags, and miniature robots. 
     Microelectronics, the development of electronic circuitry on silicon chips, is a very well developed and sophisticated technology. Micromachining utilizes process technology developed by the integrated circuit industry to fabricate tiny sensors and actuators on silicon chips. In addition to shrinking the sensor size by several orders of magnitude, integrated electronics can be placed on the same chip, creating an entire system on a chip. This instrument will result in, not only a revolution in conventional military and commercial products, but also new commercial applications that could not have existed without small, inexpensive inertial sensors. 
     MEMS inertial sensor techniques offer tremendous cost, size, reliability, power consumption improvements for guidance, navigation, and control systems, compared with conventional inertial sensors. The applicants invented a “coremicro” IMU (Inertial Measurement Unit), which is based on the combination of solid state MicroElectroMechanical Systems (MEMS) inertial sensors and Application Specific Integrated Circuits (ASIC) implementation. The “coremicro” IMU is a fully self contained motion-sensing unit. It provides angle increments, velocity increments, a time base (sync) in three axes and is capable of withstanding high vibration and acceleration. 
     The “coremicro” IMU is opening versatile commercial applications, in which conventional IMUs can not be applied. They include land navigation, automobiles, personal hand-held navigators, robotics, marine users and unmanned air users, various communication, instrumentation, guidance, navigation, and control applications. 
     The “coremicro” IMU is preferred to be employed in the present invention as the IMU  1 . But, the present invention is not limited to the “coremicro” IMU. Any IMU device with such specifications can be used in the system of the present invention. 
     Referring to FIG. 2A, the positioning assistant  8  is preferably a radio positioning system based on the wireless communication device  4 . 
     Referring to FIG. 2B, the interruption free navigator of the present invention may further comprises an altitude measurement device  100  incorporated to improve the vertical positioning performance by providing altitude measurement. There are many different altitude measurement devices, such as barometric device and radar altimeter. As shown in FIGS. 5 and 6, the main IMU based interruption-free positioning module  10  further comprises an altitude interface and processing board  91  which may include a barometric device interface and/or a radar altimeter interface for converting the altitude measurement from the altitude measurement device  100 , i.e. the barometric or the radar altimeter, into digital data of platform height over mean sea level (MSL), which is output to the integration Kalman filter  35  and further exchanged with other users through the wireless communication device  4 . 
     Referring to FIG. 2B, the interruption free navigator of the present invention may further comprises an object detection system  200 , which can be embodied as a simple and miniature camera. The object detection system  200  is used to capture images of the user&#39;s surrounding environment and to derive the position data of near objects, so that it provides a notice that an interested object has been found in the neighborhood. The capture images can be exchanged with other users through the wireless communication device  4 . The object detection system  200  may not necessarily identify the detail character of the object although it could do so on occasion. It simply alerts that there is an object in the neighborhood that merits further attention. As shown in FIGS. 5 and 6, the main IMU based interruption-free positioning module  10  further comprises an object detection system interface and processing board  92  to provide a data processing and interfacing for the object detection system  200 . 
     A conventional pure inertial navigation method is applied to a MEMS-based IMU including the “coremicro” IMU, the drift in position is too rapid to be used in an extended period. In the preferred system of the present invention, an inertial navigation system (INS) is built based on the “coremicro” IMU that is the core of the position determination system. To compensate for the error of the INS, multiple navigation sensors are integrated into the system. 
     The north finder  2  is used to measure the heading of the user. The velocity producer  6  is used to measure the relative velocity with respect to the ground. The preferred velocity producer  6  is an RF (radio frequency) radar, sonar, or a laser radar. A zero velocity updating method is used to calibrate the INS errors. The Doppler radar and zero velocity updating is used to suppress the error growth of the INS based on the established INS error model and other sensor error models. An integration Kalman filter  35 , as shown in FIG. 6, is constructed to estimate and compensate the INS errors and sensor errors. The integrated system of the present invention is used to determine the position of the user on land. 
     The preferred north finder  2  is preferably a magnetic sensor, such as a flux valve and magnetometer, sensing earth&#39;s magnetic field to measure the heading angle of the user. 
     The preferred velocity producer  6  includes an RF (radio frequency) radar, a sonar sensor, or a laser radar. 
     Based on the Doppler effect, the velocity producer  6  including a radio Doppler radar, laser Doppler, and sonar sensor can provide relative velocity measurements of the user to ground by sensing Doppler frequencies. The Doppler effect is a shift in the frequency of a wave radiated from the velocity produce when reflected by an object in motion. In the case of the present invention, Doppler shifts are produced by the relative motion of the user and the ground from which the radio or laser or sonic waves are reflected. 
     If the distance between the user and the ground is decreasing, the waves are compressed. Their wavelength is shortened and their frequency is increased. If the distance is increasing, the effect is just the opposite. The wave&#39;s Doppler frequency of the returned wave from the ground can be computed as          f   d     =     2                       v   R        cos                 L     λ                       
     where 
     ƒ d =Doppler frequency of the ground returned wave, hertz 
     V R =velocity of the radar, feet (meters) per second 
     L=angle between V R  and line of sight to the patch 
     λ=transmitted wavelength, same units as in V R . 
     Referred to FIG. 5, the positioning assistant  8  is embodied as a GPS receiver  8 A to receive interruptible GPS RF (radio frequency) signals to produce GPS raw measurements (pseudorang and range rate) or GPS position and velocity data of the user to the navigation processor  3 . If GPS signals can be available continuously, the continuous GPS raw measurements (pseudorang and range rate) or GPS position and velocity data are also incorporated in the present invention. 
     The preferred real-time software running in the navigation processor  3  further comprises the following preferred modules, as shown in FIG.  6 : 
     (3.1) an INS computation module  31 , using digital angular increments and velocity increments signals from the IMU  1  to produce inertial positioning measurements, including IMU position, velocity, and attitude data; 
     (3.2) a magnetic sensor processing module  32 , for producing the heading angle; 
     (3.3) a velocity processing module  33 , for producing relative position error measurements for a Kalman filter, and 
     (3.4) an integration Kalman filter  35 , for estimating errors of the inertial positioning measurements by means of performing, Kalman filtering computation to calibrate the inertial positioning measurement errors. 
     The IMU  1  and the associated INS computation module  31  are the core of the navigator for users on land. The INS computation module  31  further comprises: 
     a sensor compensation module  311 , for calibrating the error of the digital angular increments and velocity increments signals, which is not proportional to the user&#39;s motion; and 
     an inertial navigation algorithm module  322 , for computing the IMU position. velocity, and attitude data using the compensated the digital angular increments and velocity increments signals. 
     FIG. 9 shows the inertial navigation algorithm module  322 . While the INS provides an interruption-free, non-radiating, and deterministic means for three-dimensional navigation with accurate short-term position information, it also exhibits an unbounded position error due to uncompensated gyro and accelerometer errors, especially for low quality IMU based INS systems. The external aiding information must be provided to enhance the long-term accuracy of the system. Multiple navigation sensors in the present invention are used to aid the core INS. Flux valve aiding is used for heading updates. The velocity producer  6  and zero velocity updating are used to suppress the error growth of the INS position. Based on the established INS error model and other sensor error models, said integration Kalman filter is constructed to estimate and compensate the INS errors and sensor errors. The integrated system of the present invention is used to determine the position of the user on land 
     As the block diagram of the system of the present invention shown in FIG. 5, one of the key technologies in this present invention is the use of automatic zero-velocity updates to greatly reduce the accumulated navigation error when GPS signals are not available. The position error of an inertial navigation system (INS), which is a dead-reckoning system, increases with time with a pattern shown in FIG. 3 (A). The zero-velocity update technology uses the additional zero velocity information to reset the velocity measurement of the navigator, when the user stops. The periodic zero-velocity reset leads to a navigation error pattern shown in FIG. 3 (B). 
     With the zero-velocity reset and velocity aiding augmented with error estimation and compensation, the growth of the inertial navigation error is greatly reduced. Its navigation error pattern with time is given by the dotted line shown in FIG.  3 (B). The method of the present invention is effective to compensate so as to maintain a navigation accuracy of better than 1% of distance traveled. 
     FIG. 10 is a block diagram depicting a magnetic sensor and the magnetic sensor processing module  32 . The magnetic sensor detects the magnitude and direction of the earth&#39;s magnetic field and converts it to electrical information, which is used to obtain the magnetic north. 
     To obtain the true magnetic north, the earth&#39;s magnetic field measurement must be compensated using the measured field strengths, soft iron, and hard iron transformation matrices. 
     Ferrous metals in the user often magnetize over time, misdirecting magnetic compass readings. In addition, some devices also generate soft iron distortions. Soft iron can either misdirect or magnify existing magnetic fields, making calibration extremely difficult. 
     FIG. 11 is a block diagram depicting the velocity producer processing module  33 . 
     Referring to FIG. 6, the INS processing module  31  further comprises a sensor compensation module  311  and an inertial navigation algorithm module  312 . 
     Because the installation of the gyros and the accelerometers in an IMU is not precisely in three orthogonal directions, the IMU data must be calibrated before it is applied to the inertial navigation algorithm module  312 . 
     For example, currently, the MEMS IMU is an assembly of low accuracy MFMS silicon gyros and accelerometers. Because the MEMS sensor is very sensitive to temperature and acceleration, a set of special modules are devised in the sensor compensation module design. 
     The simplified error compensation for the MEMS gyro is expressed as: 
     
       
         ε=ε drift +ε misalign +ε nonorth +ε scale +ε g +ε random +ε t ( T ) 
       
     
     This error compensation for the MEMS IMU errors is explained as follows. 
     (1) Stability error ε drift , also denoted as drift. Since the time constant is large for the MEMS gyro stability error, this effect is modeled as a constant bias. 
     (2) Misalignment error ε misalign . Misalignment is the difference between the actual orientation of the device sensitive axis and the intended orientation of that axis. This is a constant angular error and is normally kept small by precision mounting and calibrating techniques during installation. 
     (3) Non-Orthogonal error ε nonorth . This error refers to imprecision in assembling the MEMS IMU unit itself. The IMU consists of three micro gyros which are intended to be mounted perfectly along three orthogonal axes. Non-orthogonal error results when one gyro input axis leans into the plane containing the remaining two gyro input axes. This non-orthogonal gyro will detect a component of the angular rate about the other axes. 
     (4) Scale factor error ε scale . This error is calculated as a percentage of the true angular rate sensed by the MEMS rate sensor. Scale factors produce an error in the measured angular rate of a miagnitude that is proportional to the true angular rate being measured. Scale factor errors can cause remarkable navigation errors at high angular rates. 
     (5) G sensitive error ε g . Gyro output variation due to accelerations incident on the device is called g-sensitive error. This produces a rate bias error proportional to the amount of specific force in a maneuver. The specific force is equal to the inertial acceleration of a body minus the gravitational acceleration. 
     (6) Random walk ε random . MEMS gyros are noisy sensors. In the inertial navigation system, the micro gyro acts as an integrator of the sensed angular rate. The actual sensor output of Δθ integrates the noise in the gyro output to produce a smoother signal that randomly wanders through a certain range of errors. Angle random walk is estimated by the compensation processing. 
     (7) Temperature sensitivity ε t (T). The MEMS gyro is very sensitive to a change in temperature. It can even be regarded as a good thermometer. The temperature induced error must be removed by the navigation algorithm in the INS. Thus in the IMU error processing we must provide a temperature term which can produce the error data according to the temperature changes during system operation. 
     Similarly, the error compensation for the MEMS accelerometer is expressed as: 
      ∇=∇ drift +∇ misalign +∇ nonorth +∇ scale +∇ g +∇ random +∇ t ( T ) 
     The definition of the error terms in this accelerometer are also similar to that of the MEMS gyro. 
     In practice the temperature change of an IMU can be approximately described by the first order differential equation        T   =       -     1     t   c              (     T   -     T   bal       )                       
     
       
         T(0)=T 0   
       
     
     where T represents the IMU temperature, T 0  is the initial temperature, t c  is the time constant, T bal  is the balance temperature of the IMU. Parameters t c  and T bal  are determined by the heat transfer properties of the IMU and the environmental temperature, which can be found by calibration. The temperature induced error can be expressed as 
     
       
         ε t = K   t ( T−T   nom ) 
       
     
     where T nom  is the nominal temperature at which the IMU is calibrated, and K t  is the temperature error factor. FIG. 4 shows a typical temperature induced error of a MEMS gyro. 
     Referring to FIG. 9, the inertial navigation algorithm module  322  further comprises: 
     (a) an attitude integration module  3121 , for integrating the angular increments into attitude data; 
     (b) a velocity integration module  3122 , for transforming measured velocity increments into a suitable navigation coordinate frame by using the attitude data, wherein the transformed velocity increments are integrated into velocity data; and 
     (c) a position module  3123 , for integrating the navigation frame velocity data into position data. 
     Referring to FIG. 10, the magnetic sensor processing module  32  for producing the heading angle further comprises: 
     a hard iron compensation module  321 , for receiving the digital Earth&#39;s magnetic field vector and compensating the hard iron effects in the digital earth&#39;s magnetic field vector; 
     a soft iron compensation module  322 , for compensating the soft iron effects in the digital earth&#39;s magnetic field vector; and 
     a heading computation module  322 , for receiving the compensated earth&#39;s magnetic field vector and pitch and roll from the inertial navigation algorithm module  322  and computing the heading data. 
     Referring to FIG. 11, the velocity producer processing module  33  for producing relative position error measurements for the Kalman filter module  35  further comprises: 
     a scale factor and misalignment error compensation module  331 , for compensating the scale factor and misalignment errors in the Doppler velocity; 
     a transformation module  331 , for transforming the input Doppler velocity data expressed in the body frame to the Doppler velocity expressed in the navigation frame; and 
     a relative position computation  333 , for receiving the IMU velocity and attitude data and the compensated Doppler velocity to form the relative position measurements for the Kalman filter  35 . 
     Referring to FIG.  6  and FIG. 13, the Kalman filter module  35  further comprises: 
     a motion test module  351 , for determining if the user stops automatically; 
     a GPS integrity monitor  354 , for determining if the GPS data is available; 
     a measurement and time varying matrix formation module  352 , for formulating the measurement and time varying matrix for the state estimation module  353  according to the motion status of the user from the motion test module  351  and GPS data availability from the GPS integrity monitor  354 ; and 
     a state estimation module  353 , for filtering the measurements and obtaining the optimal estimates of the IMU positioning errors. 
     In order to obtain optimal estimates of the IMU position errors, a set of error state equations need to be established for the Kalman filter. We define the relationship between the reference/ideal system determined p platform and the actual/practical system determined platform pc as 
     
       
           C   h   pc   =C   p   pc   C   h   p =( I +[φ]) C   h   p   
       
     
     where I is the identity matrix, [φ] is the skew symmetric matrix form of vector φ          [   φ   ]     =     [         0         φ   2           -     φ   y                 -     φ   z           0         φ   x               φ   y           -     φ   x           0         ]                     
     Vector φ consists of three small angle errors between the reference/ideal system determined p platform and the actual/practical system determined platform pc:        φ   =     [           φ   x               φ   y               φ   z           ]                     
     The gyro and accelerometer models are expressed as          ω   ibc   b     =       ω   ib   b     +     ɛ   b                 f   c   b     =       f   b     +     ∇   b                       
     where ε h  and ∇ h  are generalized gyro and accelerometer errors. Superscript b means the errors are expressed in the body or sensor frame. 
     Then, the generalized linear INS error model, with first order approximation, can be expressed as the following equations:          Δ                   X   .       =           ∂       F   s          (   X   )           ∂   X          Δ                 X     +     [               [   φ   ]          f   P       +     ∇   P               0         ]     +     [           Δ                   G   P               0         ]                       φ   .     =       -     ɛ   P       -       [   φ   ]          ω   ip   P       +     Δ                   ω   ip   P                     =       -     ɛ   P       -       [   φ   ]          ω   ip   P       +         ∂       F   ω          (   X   )           ∂   X          Δ                 X                             
     or in vector form 
     
       
         φ=φ×ω tp   p +Δω tp   p −ε p   
       
     
     where 
     
       
         ƒ p   =C   h   p ƒ h   
       
     
     
       
         ∇ p   =C   h   p ∇ h   
       
     
     
       
         ε p   =C   h   p ε h   
       
     
     The generalized INS error model can be used to derive specific error models for different system configurations. 
     In a preferred embodiment of the state estimation module  353 , two filters are established, namely 
     (1) a horizontal filter  3531 , for obtaining the estimates of the horizontal IMU positioning errors; and 
     (2) a vertical filter  3532 , for obtaining the estimates of the vertical IMU positioning errors. 
     The preferred state vector for the horizontal filter state vector comprises the following variables: 
     1. θ x  position error about x (rad) 
     2. θ y  position error about y (rad) 
     3. Δv x  x velocity error (F/s) 
     4. Δv y  y velocity error (F/s) 
     5. φ x  x tilt error (rad) 
     6. φ y  y tilt error (rad) 
     7. θ z  azimuth error (rad) 
     8. Δx R  relative position error about x (FT) 
     9. Δy R  relative position error about y (FT) 
     10. ε x  x gyro bias error (R/s) 
     11. ε y  y gyro bias error (R/s) 
     12. ε z  z gyro bias error (R/s) 
     13. Δθ velocity producer horizontal boresight (rad) 
     14. ΔSF velocity producer scale factor error 
     The preferred state vector for the vertical filter state vector comprises the following variables: 
     1. ΔEL=elevation error (FT) 
     2. Δv z =z velocity error (F/s) 
     3. Δz R =z relative position error (FT) 
     4. ε AZB =z accelerometer bias (F/s 2 ) 
     5. Δθ v =velocity producer vertical boresight (rad) 
     The state estimation module  353  from time to time receives the following external information: 
     (1) known position, obtained from the GPS receiver  8 A; 
     (2) known position change, obtained from the velocity processing module  33 ; 
     (3) position change equal to zero, obtained from the zero velocity update processing from the motion tests module  351 ; 
     (4) known heading, obtained from the magnetic sensor processing module  32 . 
     The difference between measured velocity from the velocity producer  6  and the IMU velocity, both resolved into a “level body” coordinate, is rapidly integrated into 3 components of relative position. 
     When the GPS data is available at every Kalman update interval, for example, every 8 seconds, the x and y relative position, x and y difference between GPS position and INS position, and x and y difference between GPS velocity and INS velocity are provided to the horizontal filter and the z relative position, z difference between GPS position and INS position, and z difference between GPS velocity and INS velocity are provided to the vertical filter. The filters will then proceed to update and correct the IMU position. 
     When the GPS data is not available at every Kalman update interval, for example, every 8 seconds, the x and y relative position are provided to the horizontal filter and the z relative position is provided to the vertical filter. The filters will then proceed to update and correct the IMU position. 
     The motion test module  351  determines if the user has stopped. If the user stops, a zero position change is applied to the Kalman filter. The definition of “stopped” is given in the following description. Suppose the Kalman update interval is 8 sec. The motion during the 8 sec period is analyzed to determine if the user “stopped” for the entire 8 sec. If so, the “stop” flag is set. To do the update with small measurement noise, the user must have been “stopped” for the proceeding 8 sec. Once the stop flag has been set, it may be reset as soon as motion is detected, which may be less than 8 sec. If the stop is reset, it may only be set by 8 continuous seconds of no motion. 
     The motion test module  351  comprises: 
     a velocity producer change test module, for receiving the velocity producer reading to determining if the user stops or restarts; 
     a system velocity change test module, for comparing system velocity change between the current interval and the previous interval to determine if the user stops or restarts; 
     a system velocity test module, for comparing the system velocity magnitude with a predetermined value to determine if the user stops or restarts: and 
     an attitude change test module, for comparing the system attitude magnitude with a predetermined value to determine if the user stops or restarts. 
     For example, in the velocity producer change test, at the start of the candidate 8 sec interval, the velocity producer input pulses are summed as they are read in. If the absolute value of the accumulation ever exceeds a pre-determined value, such as 2 pulses, at any time in the 8 sec interval, stop is reset. 
     In the system velocity change test at every pre-determined period, for example, 2 seconds, the average x, y, z velocities (as determined by x, y, z position change in 2 seconds) are compared to the previous 2-second average x, y, z velocities, respectively. If any 2-sec change in any axis exceeds a predetermined value, for example, 0.1 ft/s, the entire 8-sec interval is defined as “not stopped”. Note that a velocity of 0.1 ft/s for 2 sec corresponds to a 0.2 ft (or approximate 2 inch) of user disturbance which will be allowed. 
     In the system velocity test, if we assume an upper bound of 10 ft/s for the magnitude of system velocity error, we use these criteria: 
     Every 2 seconds compare the average x, y, z velocities with the previous 2 second average. Reset stop if any velocity is greater in magnitude than 10 ft/s. 
     In a tracked user with only one velocity producer and with the IMU mounted near the velocity producer, the user may turn by locking this track, giving no velocity producer output and small IMU velocities. This situation is detected by the attitude change test. 
     The total attitude change in any 2 second interval and in any 8 sec interval must be less than 1 degree to set the stop. 
     The GPS integrity monitor  354  receives the GPS status indication from the GPS receiver  8 A to determine if the GPS data is available. 
     Every update period of the Kalman filter (for example, 8 sec), the x, y, z relative position measurement is passed to the Kalman filter to perform an update. The scalar sigma squared=[H] [P] [H T ]+R is an estimate of the covariance of this measurement and may be used to test the reasonability of the magnitude of the measurement. Three such scalars are available, one for each axis. 
     Referring to FIG. 1, an interruption-free hand-held positioning method of the present invention comprises the following steps: 
     (1) sensing motion measurements of a user by a main IMU (Inertial Measurement Unit) to produce digital angular increments and velocity increments signals in response to a user motion, 
     (2) providing interruptible positioning data to assist said main IMU based interruption-free positioning module by a positioning assistant. 
     (3) producing interruption-free positioning data of the user using motion measurements, and improving interruption-free positioning data of the user when the interruptible positioning data is available, 
     (4) exchanging the improved interruption-free positioning data with other users by a wireless communication device, 
     (5) providing map data to obtain surrounding map information of location of the user by accessing a map database using said improved interruption-free positioning data, and 
     (6) visualizing the improved interruption-free positioning data of the user using said surrounding map information by a display device. 
     In the preferred application, the step (2) further comprises the step of: 
     2.A deducing GPS raw measurements or GPS position and velocity data by a GPS receiver; or 
     2.B deducing positioning data through a wireless communication device. 
     Referring to FIG. 5, the step 3 comprises the following steps: 
     3.1 sensing the earth&#39;s magnetic field to measure the heading angle of the user by a magnetic sensor, 
     3.2 measuring the relative velocity of the user relative to the ground by a velocity producer, and 
     3.3 blending the digital angular increments and velocity increments signals, heading angle, the relative velocity of the user relative to the ground, and the GPS raw measurements or GPS position and velocity data to produce optimal positioning data. 
     Before the step (4), the interruption-free hand-held positioning method of the present invention further comprises a step of providing altitude measurement and converting the altitude measurement into digital data of platform height over a mean sea level. 
     Before the step (4), the interruption-free hand-held positioning method of the present invention further comprises a step of capturing images of a surrounding environment of the user for deriving the position data of adjacent objects, so as to provide a notice that an interested object has been found in neighborhood. 
     In order to improve the performance, after the step (6), the interruption-free processing method of the present invention further comprises a step of: 
     (7) aiding the code and carrier phase tracking processing of the GPS signals with the velocity and acceleration data to improve the anti-jamming and high-dynamics capability of the GPS receiver. 
     The step (3.3) further comprises the following preferred modules: 
     3.3.1 computing inertial positioning measurements using the digital angular increments and velocity increments signals; 
     3.3.2 computing the heading angle using the earth&#39;s magnetic field measurements, 
     3.3.3 creating relative position error measurements in the velocity producer processing module using the relative velocity of the user relative to the ground for a Kalman filter, and 
     3.3.4 estimating errors of inertial positioning measurements by means of performing Kalman filtering computation to calibrate the inertial positioning measurements 
     In principle, step (3.3.1) can be called inertial navigation system processing. Inertial navigation is the process of calculating position by integrating velocity and computing velocity by integrating total acceleration. Total acceleration is calculated as the sum of gravitational acceleration, plus the acceleration produced by applied non-gravitational forces. In a modern INS, the attitude reference is provided by a software integration function residing in an INS computer using inputs from a three-axis set of inertial angular rate sensors. The angular rate sensor and accelerometer triad is mounted to a common rigid structure within the INS chassis to maintain precision alignment between each inertial sensor. Such an arrangement has been denoted as a strapdown INS because of the rigid attachment of the inertial sensors whittling the chassis, hence, to the user in which the INS is mounted. 
     The primary functions executed in the INS computation module  31  are the angular rate integration into attitude function, denoted as attitude integration, use of the attitude data to transform the measured acceleration into a suitable navigation coordinate frame where it is integrated into velocity, denoted as velocity integration, and integration of navigation frame velocity into position, denoted as position integration. Thus, three integration functions are involved, attitude, velocity, and position, each of which requires high accuracy to assure negligible error compared to inertial sensor accuracy requirements. 
     Therefore, step (3.3.1) further comprises the steps of: 
     3.3.1.1 integrating the angular increments into attitude data, referred to as attitude integration processing; 
     3.3.1.2 transforming the measured velocity increments into a suitable navigation coordinate frame by use of the attitude data, wherein the transferred velocity increments are integrated into velocity data, denoted as velocity integration processing, and 
     3.3.1.3 integrating the navigation frame velocity data into position data, denoted as position integration processing. 
     In the strapdown INS, a mathematical frame (or an imaginary frame) is introduced, which emulates the motion of a level platform, so it is also called the platform frame and denoted by P. The user velocity relative to the earth is represented in this mathematical frame. Written in the compact vector form, the velocity integration equation of the strapdown INS can be expressed as follows:          V   .     =       f   P     +     G   P     -       (       ω   cp   P     +     2                   Ω   ic   P         )     ×   V                       
     where 
     V is the user velocity relative to the earth, represented in the P frame. 
     ƒ p  is the specific force represented in the P frame, or the accelerometer output transformed to the mathematical platform. 
     G p  is the gravitational acceleration represented in the P frame. 
     ω ep   p  is the angular velocity of the mathematical platform with respect to the earth frame expressed in the P frame. 
     Ω tc   p  s the earth rate represented in the P frame. 
     In order to obtain a definite velocity equation for the INS, we have to first define the motion of the mathematical platform. 
     The P frame in the strapdown INS is a level platform, so its angular position with respect to the local geographical frame (N frame) can be described by an azimuth angle α. The angular velocity of the platform with respect to the inertial frame can be expressed as:          ω   ip   P     =       ω   np   P     +       C   n   P          ω   en   n       +       C   n   P          Ω   ie   P                         
     where 
     ω np   p  is the angular velocity of the P frame with respect to the N frame. 
     C n   p  is direction cosine matrix of the P frame relative to the N frame. 
     ω en   n  is the angular velocity of the local geographical frame (N frame) with respect to the earth frame (E frame). 
     Since the P frame is a mathematical platform, we are able to define its motion. Based on the above equation, we can obtain an equation to describe the motion of the P frame relative to the N frame:          ω   ipz   P     =       α   .     +         tg                 ϕ         R   n     +   h            v   x   n       +     Ω                 sin                 ϕ                       
     We define ω ipz   p  to obtain different mechanization for the INS. In analogy to the gimbaled INS, we let ω ipz   p =0 to have a so-called free-azimuth system, or let ω ipz   p ΩsinΦ have a so-called wander-azimuth system. Thus we have the motion of the mathematical platform defined. 
     For the free-azimuth system            α   .     =         -       tg                 ϕ         R   n     +   h              v   x   n       -     Ω                 sin                 ϕ                                    
     For the wander-azimuth system          α   .     =       -       tg                 ϕ         R   n     +   h              v   x   n                       
     As long as the motion of the P frame is defined, we arrive at a definite velocity equation for the strapdown INS. Further, we can obtain a third-order, nonlinear, time-varying, ordinary differential equation as the INS velocity equation. 
     Expressed in geographical latitude and longitude, the position integration equation of the INS is written as          ϕ   .     =         v   y   n         R   m     +   h       =       1       R   m     +   h            (         v   x        sin                 α     +       v   y        cos                 α       )                   λ     ^   .       =         v   x   n         (       R   n     +   h     )        cos                 ϕ       =       1       (       R   n     +   h     )        cos                 ϕ            (         v   x        cos                 α     -       v   y        sin                 α       )                         
     It is noted that the longitude equation has a singularity at the earth&#39;s two poles. The longitude computation will become difficult in the polar areas. In practice, if the polar area navigation is required, we can introduce other position representation variables. For example, we can use the direction cosine matrix of the N frame relative to the earth centered earth fixed frame (ECEF frame), C e   n , as the position variable. Then the position equation is expressed as:          C   c   n     =       [     ω   en   n     ]          C   e   n                       
     This is a matrix differential equation. Where [ω en   n ] is the skew-symmetric matrix corresponding to the vector ω en   n . This differential equation has no singular point and from C e   n , we can calculate the position represented in latitude and longitude. In this equation, however, C e   n  has 9 elements but 3 degrees-of-freedom. Thus in the computation, a normalization procedure is required to keep the C e   n  normalized. That is, at every integration step, we modify C e   n , making it satisfy 
     
       
           C   e   n   C   e   n1 =1 
       
     
     If we regard the INS velocity equation as a nonlinear, time-varying system, the specific force in the P frame, ƒ p , and the gravitational acceleration, G p , can be regarded as the inputs to the system. If we ignore the gravity anomaly, the gravitational acceleration can be represented as          G   p     =     [         0           0           g         ]                     
     where g is the normal gravity that can be expressed as 
     
       
           g=g   0 [1−2 A ( {fraction (h/α)})+   B  sin 2 Φ] 
       
     
     where 
     A=1+ƒ+m 
       B =2.5 m −ƒ 
     ƒ=flattening of the reference ellipsoid. 
       m=Ω   2 α 2   b/GM    
       g   0 =equatorial gravity. 
       h =altitude 
       M =mass of the earth. 
       G =gravitational constant. 
     The specific force in the P frame, ƒ p , is the actual accelerometer output transformed into the mathematical platform: 
     
       
         ƒ p   =C   h   p ƒ h   
       
     
     where ƒ h  is the accelerometer output vector or the specific force represented in the IMU frame (or the body frame). To carry out this transformation, the direction cosine matrix C h   n  must be known. That is, the attitude of the IMU frame must be obtained. In the strapdown INS the attitude is obtained from high-speed computation. It is through the attitude computation and coordinate system transformation, that the mathematical platform is established. In the implementation of the strapdown INS, the attitude computation is the most critical issue. 
     In principle, there are many kinds of parameters used to represent the attitude of a rigid body. For example, Euler angles, direction cosine matrix, quaternion, Euler parameters, etc. In practice, the direction cosine matrix and the quaternion are the most frequently used attitude representations in the analysis and computations. Represented with the direction cosine matrix, the attitude differential equation is written as:            C   .     p   b     =       [     ω   pb   b     ]          C   p   b                       
     This is a matrix differential equation.        [     ω   pb   b     ]                   
     is the skew-symmetric matrix corresponding to the vector ω ph   h  that is determined by the following equation:                ω   pb   b     =                  ω   ib   b     -     ω   ip   b                   =                  ω   ib   b     -       C   p   b          (       ω   ep   p     +       C   n   p          Ω   ie   n         )                     =                  ω   ib   b     -       C   p   b          ω   ip   p                               
     where ω th   h  is the output of the gyros in the IMU or the IMU angular velocity with respect to the inertial space represented in the IMU frame itself. 
     We can see that the attitude equation is a 9 th -order, nonlinear, time-varying ordinary differential equation. In this equation, however, the C p   h  has 9 elements but 3 degrees-of-freedom. Thus, in the computation, a normalization procedure is required to keep the C p   h  normalized. That is, in every integration step, we modify C p   h , making it satisfy            C   p   b          C   p     b   T         =     I   .                     
     The quaternion representation is often used in the attitude computation because of its conciseness and efficiency. The quaternion attitude equation is expressed as:          λ   .     =       1   2        ωλ                     
     where λ is the quaternion expressed in the column matrix form, and ω is the 4×4 matrix determined by the angular velocity ω ph   h .        λ   =     [           λ   0               λ   1               λ   2               λ   3           ]             ω   =     [         0         -     ω   x             -     ω   y             -     ω   z                 ω   x         0         ω   z           -     ω   y                 ω   y           -     ω   z           0         ω   x               ω   z           ω   y           -     ω   x           0         ]                     
     The quaternion has 4 parameters to represent the body attitude, while a rigid body has only 3 degrees-of-freedom. Thus the components of the quaternion are constrained by the relationship:            λ   0   2     +     λ   1   2     +     λ   2   2     +     λ   3   2       =   1                   
     The quaternion satisfying this equation is called normalized. In the integration of the attitude equation, the normalization of the quaternion is also very simple. The relation between the quaternion and the direction cosine matrix is expressed as follows:          C   p   b     =         [           λ   1               λ   2               λ   3           ]                [           λ   1           λ   2           λ   3           ]     +       [           λ   0           λ   3           -     λ   2                 -     λ   3             λ   0           λ   0               λ   2           -     λ   0             λ   0           ]     2                       
       
     To express the INS model in a compact form we introduce a vector defined as:        X   =       [         V           α           ϕ           λ         ]     =     [           v   x               v   y               v   z             α           ϕ           λ         ]                       
     Then the INS computation model can be written as                X   .     =         F   x          (   X   )       +     [             C   b   p          f   h               O         ]     +     [           G   p             O         ]                     ω   ip   p     =       F   ω          (   X   )                       C   .     p   b     =       [     ω   pb   b     ]          C   p   b                     ω   pb   b     =       ω   ib   b     -       C   p   b          ω   ip   p                               
     Referring to FIG. 13, the step (3.3.4) further comprises the steps of: 
     3.3.4.1 performing motion tests to determine if the user stops to initiate the zero-velocity update; 
     3.3.4.2 determining if GPS data available using the GPS state status indicator from the GPS receiver; 
     3.3.4.3 formulating measurement equations and time varying matrix for the Kalman filter; and 
     3.3.4.4 computing estimates of the error states using Kalman filter. 
     A flow diagram of the preferred implementation in step (3.2) to form the measurement is given in FIG.  12 . 
     The variables and parameters in FIG. 10 are defined and described as follows: 
     ΔD, Doppler frequency pulses. 
     ΔT, time interval of high speed navigation/velocity producer loop. 
     ΔD/ΔT, velocity indicated by the velocity producer. 
     SFC, scale factor in pulse/(F/s). 
     V ODC.  computed velocity. 
     Δθ 1  and Δθ 3.  horizontal and vertical velocity producer boresight estimates. 
     PIT and ROL, IMU pitch, roll. 
     V x,y,z.  navigation system (wander α) velocity. 
     AAC, sum of computer α and computer heading. The transformations using PIT, ROL, AAC must be current at high speed navigation rate. 
     Referring to FIG. 12, the step (3.2) further comprises: 
     (3.2.1) transforming the measured velocity expressed in the body frame into the navigation frame; 
     (3.2.2) comparing the measured velocity with the IMU velocity to form a velocity difference; and 
     (3.2.3) integrating the velocity difference during the predetermined interval. 
     In the preferred application, the step (2) can be disclosed as: 
     2C. deducing differential GPS positioning data through a GPS receiver and a data link. 
     It is well known that the receiver measurement noise for the L 1  and L 2  frequencies is about 1.9 mm and 2.4 mm, respectively, while the receiver measurement noise for P(Y) and C/A codes is about 0.3 m and 3 m, respectively. Therefore, the above embodiment of the present patent is enhanced with the differential GPS (DGPS) carrier phase method. 
     Referring to FIG. 7, in order to exploit DGPS, the positioning assistant  8  further comprises: 
     a GPS receiver  8 A, for receiving GPS RF (radio frequency) signals to produce GPS raw measurements (pseudorange, range rate, and carrier phase) or GPS position and velocity data; and 
     a data link  8 B, for receiving the position, velocity, and raw measurements (pseudorange and phase) from a GPS reference site to perform differential GPS positioning; 
     In order to improve the performance, velocity and acceleration data from the navigation processor are fed back to the GPS receiver to aid the code and carrier phase tracking of the GPS signals. 
     However, the high accuracy of positioning with GPS carrier phase measurements is based on the prior condition that the phase ambiguities have been resolved. &#39;The ambiguity inherent with phase measurements depends upon both the GPS receiver and the satellite. Under the ideal assumptions of no carrier phase tracking error and the known true locations of the receiver and satellite, the ambiguity can be resolved instantaneously through a simple math computation. However, there is the presence of satellite ephemeris error, GPS satellite clock bias, atmospheric propagation delay, multipath effect, receiver clock error and receiver noise in range measurements from the GPS code and phase tracking loops. Therefore, a phase ambiguities resolution is disclosed here to apply DGPS to the above embodiment. 
     The GPS receiver  8 A receives GPS RF (radio frequency) signals from the GPS satellites and outputs the pseudorange. Doppler shifts. GPS satellite ephemerides, as well as atmospheric parameters to the Kalman filter  35 . 
     Correspondingly, referring to FIG. 8, the preferred real-time software running in the navigation processor  3  further comprises: 
     a new satellites/cycle slips detection module  36 , receiving the GPS measurements from the GPS receiver  8 A and GPS reference measurement from the data link  8 B and determines if new GPS satellites come in view or cycle slips occur; and 
     an on-the-fly ambiguity resolution module  37 , receiving the GPS measurements from the GPS receiver  8 A and GPS reference measurement from the data link  8 B and is activated when new GPS satellites come in view or cycle slips occur to fix the ambiguity integer. 
     For GPS measurements, the double difference equations for L 1  and L 2  frequencies are (scalar equations)                P     k                 mr     ij     =                  ρ   mr   ij     +     ρ     c                 mr     ij     +       I   mr   ij       f   k   2       +     T   mr   ij     +     d     pc     k                 mr       ij     +     M     P     k                 mr       ij     +     ɛ     P     k                 mr       ij                     Φ     k                 mr     ij     =                  ρ   mr   ij     +     ρ     c                 mr     ij     +       λ   k          N     k                 mr     ij       -       I   mr   ij       f   k   2       +     T   mr   ij     +     d     pc     k                 mr       ij     +                                  M     Φ                 c                 mr     ij     +     ɛ     Φ     k                 mr       ij       ,     (       k   =   1     ,   2     )                             
     where (.) mr   y  means double difference which is formed by (.) m   i −(.) m   j −(.) r   i +(.) r   j . The subscripts m and r denote two (reference and rover) receivers and the superscripts i and j represent two different GPS satellites. P and Φ are the pseudorange and phase range measurements, respectively, ρ is the geometric distance between the phase centers of two antennas (a GPS user&#39;s receiver and a GPS satellite) at the nominal time and ρ refers to the correction of nominal geometrical distance, λ represents wavelength. N mr   y  is the double difference integer ambiguity.          I   mr   ij       f   k   2                     
     is the double difference residual of the ionospheric effect for L 1  or L 2  frequency and T mr   y  denotes the double difference residual of the tropospheric effect, d pc(.)     mr     y  refers as the double difference residuals of phase center variations. M (.)     mr     y  denotes the double difference residuals of multipath effect. The definitions of the wide lane and narrow lane phase range measurements are                  Φ     w                 mr     ij     =           f   1         f   1     -     f   2              Φ     1                 mr     ij       -         f   2         f   1     -     f   2              Φ     2      mr     ij                       Φ     n                 mr     ij     =           f   1         f   1     +     f   2              Φ     1                 mr     ij       +         f   2         f   1     +     f   2              Φ     2      mr     ij                 ,                   
     respectively, and the corresponding integer ambiguities are                  N     w                 mr     ij     =       N     1      mr     ij     -     N     2      mr     ij                     N     n                 mr     ij     =       N     1      mr     ij     +     N     2      mr     ij               ,                   
     respectively. Therefore, the frequencies for the wide lane and narrow lane ambiguities are equal to  1 ƒ w =ƒ 1 −ƒ 2  and ƒ n =ƒ 1 +ƒ 2 , respectively. Linearly combining the L 1  and L 2  equations and using t k  to represent time at epoch k, the sequentially averaged approximated double difference wide lane ambiguity (real number) is expressed as                    N   _       w                 mr     ij          (     t   k     )       =           ∑     i   =   1     k                         N   ~       w                 mr     ij          (     t   i     )         k     =           (     k   -   1     )              N   _       w                 mr     ij          (     t     k   -   1       )         +         N   ~       w                 mr     ij          (     t   k     )         k               (   1   )                         
     and the approximated double difference narrow lane ambiguity (real number) is given by                    N   ~       n                 mr     ij     ≈       (         λ   w          N     w                 mr     ij       -       Φ   _       IS                 mr     ij     +     d     pc     w                 mr       ij     -     d     pc     n                 mr       ij       )     /     λ   n         ,           (   2   )                   where                     N   ~       w                 mr     ij       ≈       (       Φ     w                 mr     ij     -     P     n                 mr     ij     -     d     pc     n                 mr       ij     +     d     pc     n                 mr       ij       )     /     λ   w         ,                                   Φ   _       IS                 mr     ij          (     t   k     )       =           ∑     i   =   1     k                     Φ     IS                 mr     ij       k     =           (     k   -   1     )              Φ   _       IS                 mr     ij          (     t     k   -   1       )         +       Φ     IS                 mr     ij          (     t   k     )         k         ,                                     
     Φ ISmr   ij =Φ nmr   ij −Φ wmr   ij  denotes the ionospheric signal observation,            P     n                 mr     ij     =           f   1         f   1     +     f   2              P     1      mr     ij       +         f   2         f   1     +     f   2              P     2                 mr     ij           ,                             d     pc     w                 mr       ij     =           f   1         f   1     -     f   2              d     pc     1                 mr       ij       -         f   2         f   1     -     f   2              d     pc     2                 mr       ij           ,   and             d     pc     n                 mr       ij     =           f   1         f   1     -     f   2              d     pc     1                 mr       ij       +         f   2         f   1     +     f   2                d     pc     2                 mr       ij     ·     λ   w                           
     and λ n  refer to the wavelengths of the wide lane and narrow lane ambiguities, respectively. Also, the ionosphere-free models for pseudorange and phase range are defined as                  P     IP                 mr     ij     =           f   1   2         f   1   2     -     f   2   2              P     1                 mr     ij       -         f   2   2         f   1   2     -     f   2   2              P     2                 mr     ij                       Φ     Ib                 mr     ij     =           f   1   2         f   1   2     -     f   2   2              Φ     1                 mr     ij       -         f   2   2         f   1   2     -     f   2   2              Φ     2                 mr     ij                 ,                   
     respectively. 
     Referring to FIG. 8, the on-the-fly ambiguity resolution module  37  is activated when the new satellites/cycle slips detection module  36  is on. And, therefore, the rover raw and Doppler shift measurements from the GPS receiver  8 A and the reference raw measurements, Doppler shift measurements, position, and velocity from the data link  8 B to fix the integer ambiguities. After fixing of the integer ambiguities, the integer ambiguities are passed to the Kalman filter  35  to further improve the measurement accuracy of the GPS raw data. 
     FIGS. 14,  15 ,  16 , and  17  represent the method and process used for the on-the-fly ambiguity resolution module  37 . FIG. 14 shows the process of the on-the-fly ambiguity resolution module  37 . When the on-the-fly ambiguity resolution module  37  is on, a search window is set up. The search window comprises several time epochs (assumed N epochs). Within the search window, an intermediate ambiguity search strategy (IASS) is used to search an integer ambiguity set at each epoch. 
     The on-the-fly ambiguity resolution module  37  performs the following steps: 
     (a) initiating an on-the-fly ambiguity resolution module as the new satellites/cycle slips detection module is on, i.e., the new satellites or cycle slips occur; 
     (b) fixing integer ambiguities to estimate a more accurate user navigation solution, and 
     (c) sending the selected integer ambiguities from the on-the-fly ambiguity resolution module  37  to the Kalman filter  35 . 
     The above step (b) further comprises the steps of: 
     (b.1) using intermediate ambiguity search strategy (IASS) and estimator bank to set up ambiguity set and determine the ambiguity integer; and 
     (b.2) validating and confirming the ambiguity integer. 
     Basically, IASS comprises the “simplified” least-squares method and the extrawidelaning technique. Before using the least-squares method to search the ambiguities, the observable common satellites between two antennas (reference and rover) are divided into two groups: 
     (1) the primary satellites. Since the double difference equations are used, the satellite with the highest elevation is defined as the reference satellite. The primary satellites include the next four higher elevation satellites. i.e., there are four independent double difference equations. 
     (2) the secondary satellites. The rest of the observable satellites are categorized into the secondary satellites. 
     As shown in FIG. 15, the IASS process comprises of a primary double difference wide lane ambiguity resolution module  371 , an ambiguity domain determination module  372 , a least-squares search estimator  373 , a position calculation module  374 , a secondary double difference wide lane ambiguity resolution module  375 , an extrawidelaning technique module  376 , and an L 1  and L 2  ambiguity resolution module  377 . 
     The first step of the IASS is to resolve the primary double difference wide lane ambiguities in the primary double difference wide lane ambiguity resolution module. The a priori information about the rover position (obtained from ionosphere-free pseudorange measurements) and the approximated double difference wide lane ambiguities (Equation (1)) are combined with the primary double difference wide lane phase range measurements to form the simultaneous equations. Also, the a priori information about the rover position can be given by the output of the navigation processor  3 . Use the minimum variance with a priori information to estimate the rover position and primary double difference wide lane ambiguities. 
     After the estimation of the primary double difference wide lane ambiguities, the estimated primary double difference wide lane ambiguities and the corresponding, cofactor matrix are sent to the ambiguity domain determination module, wherein an ambiguity search domain is established based on the estimated double difference wide lane ambiguities and the corresponding cofactor matrix. The ambiguity search domain is sent to the least-squares search estimator. A standard least-squares search method is applied to search the ambiguity set in the least-squares search estimator. Also, the standard least-squares search method can be simplified to accelerate the ambiguity search. The “simplified” least-squares search method is defined as directly searching the ambiguity set, that minimizes the quadratic form of the residuals 
     
       
           R   1 =( {circumflex over (x)}   N   −n   1 ) T   P   {circumflex over (x)}     N     −1 ( {circumflex over (x)}   N   −n   i ) 
       
     
     where {circumflex over (x)} N  is the optimal estimate vector of the double difference wide lane ambiguities (real number), n i  is the double difference wide lane ambiguity vector within the search domain (integer number), and P {circumflex over (x)}     N    is the cofactor matrix corresponding, to the optimal double difference wide lane ambiguity estimate, without using statistical or empirical tests (because the estimator bank will execute the task of confirmation). 
     The fixed primary double difference wide lane ambiguities are sent to the position calculation module to compute the rover position. The calculated rover position is sent to the secondary double difference wide lane ambiguity resolution module to fix the secondary double difference wide lane ambiguities by applying the primary wide-lane-ambiguity-fixed rover position solution into the secondary double difference wide lane phase measurements. 
     Substituting the resolved double difference wide lane ambiguities into Equation (2), the approximated double difference narrow lane ambiguities (real numbers) are calculated. The extrawidelaning technique states that if the wide lane ambiguity is even (or odd), the corresponding narrow ambiguity is even (or odd), and vice versa. Using the extrawidelaning technique, the narrow lane ambiguities can be resolved in the extrawidelaning technique module. Therefore, in the L 1  and L 2  ambiguity resolution module, the L 1  and L 2  ambiguities are calculated from the combination of the wide lane ambiguities and narrow lane ambiguities, which correspond to          N     1                 mr     ij     =         N     w                 mr     ij     +     N     n                 mr     ij       2                     
     and            N     2                 mr     ij     =         N     n                 mr     ij     -     N     w                 mr     ij       2       ,                   
     respectively. 
     Returning to FIG. 14, when the current ambiguity set from the IASS is different from the one(s) from the previous epoch(s), the current ambiguity set becomes a new member of an estimator bank  378  and a corresponding weight bank  379 . When the current ambiguity set is the same as one of the previous ambiguity set(s) in the estimator bank  378 , the number of Kalman filters in the estimator bank  378  remains. The complete form of the estimator bank  378  and weight bank  379  is depicted in FIG.  17 . The process for establishing the estimator bank and weight bank is shown in FIG.  16  and comprises the following steps: 
     Search the integer ambiguity set at the first epoch of the search window by using the IASS. The integer ambiguity set becomes a member of the estimator bank because there is no member in the estimator bank  378  before the first epoch. Based on the ambiguity set and phase measurements, the rover position (ambiguity-fixed solution) is estimated, and then a corresponding weight is calculated in the weight bank. The calculation of the weight is according to                    p   m          (       N   i     |     ΔΦ   k   *       )       =         p   m          (       ΔΦ   k   *     |     N   i       )           ∑     i   =   1     D                       p   m          (       ΔΦ   k   *     |     N   i       )             ,     i   =   1     ,   2   ,   …              ,   D           (   3   )                         
     where 
     
       
           p   m (ΔΦ k   *|N   1 )= p   m (ΔΦ k |ΔΦ k−1 , ΔΦ k−2 . . . , ΔΦ 1   , N   1 ) p   m (ΔΦ k−1   *|N   1 ),  i =1,2 , . . . D   (4) 
       
     
     and the first term of the product can be expressed as              p   m          (         ΔΦ   k          ΔΦ     k   -   1         ,     ΔΦ     k   -   2       ,     …                   ΔΦ   1       ,     N   1       )       =       1           (     2      π     )     ′          det        (     cov        (     ΔΦ   k     )       )                  exp        (     -           z   ^     k   T            (     cov        (     ΔΦ   k     )       )       -   1              z   ^     k       2       )           ,     i   =   1     ,   2   ,     …                 D                     
     which is assumed and defined as a Gaussian distribution. Equation (4) states the accumulative property of p m (ΔΦ k *|N 1 ), where p m (ΔΦ k *|N 1 ) represents the probability mass function of the measurement sequence ΔΦ k *={ΔΦ 1 , ΔΦ 2 , . . . , ΔΦ k } up to the current time t k  conditioned by the individual ambiguity set N 1 . In other words, the calculation of the weight depends on not only the data of the current epoch but also the data of the previous epochs, det(.) and (.) −1  denote the determinant and the inverse of a matrix, respectively, {circumflex over (z)} k  is the optimal measurement residual (measurement value—the optimal computed value) at time t k  and cov(ΔΦ k )=E{{circumflex over (z)} k {circumflex over (z)} k   T } is the covariance matrix of the measurement at the time t k , r is the dimension of the measurement at each epoch. For the first epoch (t 1 ) (k=1) of the search window, Equation (4) (probability) becomes                    p   m          (       ΔΦ   1   *          N   t       )       =       1           (     2      π     )     ′          det        (     cov        (     ΔΦ   k     )       )             ·     exp        (     -           z   ^     k   T            cov        (     ΔΦ   k     )         -   1              z   ^     k       2       )           ,     i   =   1     ,   2   ,   …              ,     D   .             (   5   )                         
     Of course, the value of the only weight (D=1 in Equation (3)) in the weight bank  379  is equal to one. The optimal rover position for this epoch is equal to the rover position multiplied by the corresponding weight. Based on the optimal rover position and the Doppler shifts, the rover velocity is estimated. 
     Search the ambiguity set by using the IASS at the second epoch of the search window. Two situations may occur: 
     2-1. When the integer ambiguity set is the same as the one of the previous epoch (epoch one), the number of the Kalman filters in the estimator bank  378  is still one, as shown in the lower part of FIG.  16 . Based on the ambiguity set and the phase measurements (for epoch two), the rover position (ambiguity-fixed solution) can be estimated and the corresponding weight in the weight bank is calculated cumulatively (i.e. Equations (3) and (4), where D=1). The optimal rover position for epoch two is equal to the rover position multiplied by the associated weight (naturally, for this case the value of the weight is equal to one). Based on the optimal rover position and the Doppler shifts, the rover velocity is estimated. 
     2-2. When the integer ambiguity set is different from the one of the previous epoch (epoch one), the current ambiguity set becomes a new member of the estimator bank, i.e., the number of the Kalman filters in the estimator bank  378  is two, as shown in the upper part of FIG.  16 . Based on each ambiguity set and the same phase measurements (for epoch two), the individual rover position (ambiguity-fixed solution) can be estimated and the calculation of each corresponding weight in the weight bank is based on Equations (3) and (5) (where D=2). In other words, when new ambiguity set is resolved, each corresponding weight in the weight bank  379  is calculated from scratch. The optimal rover position for epoch two is equal to the sum of the individual rover position multiplied by the associated weight. Based on the optimal rover position and the Doppler shifts, the rover velocity is estimated. 
     Follow the same procedure as described in step 2 for the rest of the epochs of the search window. At the last epoch (epoch N) of the search window, after the search of IASS, the estimator bank  378  and weight bank  379  are completely established (referred to FIG.  17 ). 
     Referring to FIG. 14, after the period of the search window, still, the phase measurements (reference and rover) are input into the complete estimator bank  378  (as shown in FIG.  17 ). As shown in FIG. 17, each Kalman filter in the estimator bank  378  has its own ambiguity set, which is selected by the IASS during the search window. Therefore, the number of the Kalman filters, D, in the estimator bank  378  is an arbitrary positive integer number which depends on the number of the different ambiguity sets from the search of the IASS during the search window. Based on each ambiguity set and the phase measurements, the individual rover position (ambiguity-fixed solution) can be estimated and each corresponding weight in the weight bank is calculated cumulatively (based on Equations (3) and (4)). Thus, the optimal rover position is equal to the sum of the individual rover position multiplied by the associated weight. Based on the optimal rover position and the Doppler shifts, the rover velocity is estimated. Follow the same procedure until a criterion is met. The criterion is defined as 
     
       
           p   m (ΔΦ k   *|N   1 )&gt; C   
       
     
     where C denotes a very large uncertainty to make sure that the ambiguity set is robust enough. After the criterion is met, the estimator bank  378  and weight bank  379  stop functioning and output the selected integer ambiguities into the Kalman filter (referred to FIG.  17 ). During the confirmation period (from the first epoch of the search window to the epoch when the estimator bank  378  and weight bank  379  stop functioning), the estimator bank  378  and the weight bank  379  identify the correct integer ambiguity set and estimates the rover position in real time. One important characteristic of the estimator bank  378  and weight bank  379  is that the weight in the weight bank corresponding to the collect integer ambiguity in the estimator bank  378  is approaching one while the others (corresponding to the rest of the ambiguity sets) are converging to zero. Therefore, the correct (selected) integer ambiguity set is the one with the weight close to one. During the whole procedure, when new satellites or cycle slips occur, the on-the-fly ambiguity resolution module  37  will be initiated (on). 
     According to the preferred embodiment of the present invention, the preferred IMU  1  is embodied with the coremicro MEMS IMU in which a position and attitude processor is built in. The position and attitude processor can replace the above disclosed INS computation module  31 . The coremicro IMU is disclosed as follows. 
     Generally, an inertial measurement unit (IMU) is employed to determine the motion of a carrier. In principle, an inertial measurement unit relies on three orthogonally mounted inertial angular rate producers and three orthogonally mounted acceleration producers to obtain three-axis angular rate and acceleration measurement signals. The three orthogonally mounted inertial angular rate producers and three orthogonally mounted acceleration producers with additional supporting mechanical structure and electronic devices are conventionally called an Inertial Measurement Unit (IMU). The conventional IMUs may be cataloged into Platform IMU and Strapdown IMU. 
     In the platform IMU, angular rate producers and acceleration producers are installed on a stabilized platform. Attitude measurements can be directly picked off from the platform structure. But attitude rate measurements can not be directly obtained from the platform. Moreover, there are highly accurate feedback control loops associated with the platform. 
     Compared with the platform IMU, in the strapdown IMU, angular rate producers and acceleration producers are directly strapped down with the carrier and move with the carrier. The output signals of the strapdown rate producers and acceleration producers re expressed in the carrier body frame. The attitude and attitude rate measurements can be obtained by means of a series of computations. 
     A conventional IMU uses a variety of inertial angular rate producers and acceleration producers. Conventional inertial angular rate producers include iron spinning wheel gyros and optical gyros, such as Floated Integrating Gyros (FIG), Dynamically Tuned Gyros (DTG), Ring Laser Gyros (RLG), Fiber-Optic Gyros (FOG). Electrostatic Gyros (ESG), Josephson Junction Gyros (JJG), Hemisperical Resonating Gyros (HRG), etc. Conventional acceleration producers include Pulsed Integrating Pendulous Accelerometer (PIPA), Pendulous Integrating Gyro Accelerometer (PIGA), etc. 
     The processing method, mechanical supporting structures, and electronic circuitry of conventional IMUs vary with the type of gyros and accelerometers employed in the IMUs. Because conventional gyros and accelerometers have a large size, high power consumption, and moving mass, complex feedback control loops are required to obtain stable motion measurements. For example, dynamic-tuned gyros and accelerometers need force-rebalance loops to create a moving mass idle position. There are often pulse modulation force-rebalance circuits associated with dynamic-tuned gyros and accelerometer based IMUs. Therefore, conventional IMUs commonly have the following features: 
     High cost, 
     Large bulk (volume, mass, large weight), 
     High power consumption, 
     Limited lifetime, and 
     Long turn-on time. 
     These present deficiencies of conventional IMUs prohibit them from use in the emerging commercial applications, such as phased array antennas for mobile communications, automotive navigation, and hand-held equipment. 
     New horizons are opening up for inertial sensor device technologies. MEMS (MicroElectronicMechanicalSystem) inertial sensors offer tremendous cost, size, and reliability improvements for guidance, navigation, and control systems, compared with conventional inertial sensors. 
     MEMS, or, as stated more simply, micromachines, are considered as the next logical step in the silicon revolution. It is believed that this coming step will be different, and more important than simply packing more transistors onto silicon. The hallmark of the next thirty years of the silicon revolution will be the incorporation of new types of functionality onto the chip structures, which will enable the chip to, not only think, but to sense, act, and communicate as well. 
     Prolific MEMS angular rate sensor approaches have been developed to meet the need for inexpensive yet reliable angular rate sensors in fields ranging from automotive to consumer electronics. Single input axis MEMS angular rate sensors are based on either translational resonance, such as tuning forks, or structural mode resonance, such as vibrating rings. Moreover, dual input axis MEMS angular rate sensors may be based on angular resonance of a rotating rigid rotor suspended by torsional springs. Current MEMS angular rate sensors are primarily based on an electronically-driven tuning fork method. 
     More accurate MEMS accelerometers are the force rebalance type that use closed-loop capacitive sensing and electrostatic forcing. Draper&#39;s micromechnical accelerometer is a typical example, where the accelerometer is a monolithic silicon structure consisting of a torsional pendulum with capacitive readout and electrostatic torque. Analog Device&#39;s MEMS accelerometer has an integrated polysilicon capacitive structure fabricated with on-chip BiMOS process to include a precision voltage reference, local oscillators, amplifiers, demodulators, force rebalance loop and self-test functions. 
     Although the MEMS angular rate sensors and MEMS accelerometers are available commercially and have achieved micro chip-size and low power consumption, however, there is not yet available high performance, small size, and low power consumption IMUs. 
     Currently, MEMS exploits the existing microelectronics infrastructure to create complex machines with micron feature sizes. These machines can have many functions, including sensing, communication, and actuation. Extensive applications for these devices exist in a wide variety of commercial systems. 
     The difficulties for building a coremicro IMU is the achievement of the following hallmark using existing low cost and low accuracy angular rate sensors and accelerometers: 
     Low cost, 
     Micro size 
     Lightweight 
     Low power consumption 
     No wear/extended lifetime 
     Instant turn-on 
     Large dynamic range 
     High sensitivity 
     High stability 
     High accuracy 
     To achieve the high degree of performance mentioned above, a number of problems need to be addressed: 
     (1) Micro-size angular rate sensors and accelerometers need to be obtained. Currently, the best candidate angular rate sensor and accelerometer to meet the micro size are MEMS angular rate sensors and MEMS accelerometers. 
     (2) Associated mechanical structures need to be designed. 
     (3) Associated electronic circuitry needs to be designed. 
     (4) Associated thermal requirements design need to be met to compensate the MEMS sensor&#39;s thermal effects. 
     (5) The size and power of the associated electronic circuitry needs to be reduced. 
     The coremicro inertial measurement unit of the present invention is preferred to employ with the angular rate producer, such as MEMS angular rate device array or gyro array, that provides three-axis angular rate measurement signals of a carrier, and the acceleration producer, such as MEMS acceleration device array or accelerometer array, that provides three-axis acceleration measurement signals of the carrier, wherein the motion measurements of the carrier, such as attitude and heading angles, are achieved by means of processing procedures of the three-axis angular rate measurement signals from the angular rate producer and the three-axis acceleration measurement signals from the acceleration producer. 
     In the present invention, output signals of the angular rate producer and acceleration producer are processed to obtain digital highly accurate angular rate increment and velocity increment measurements of the carrier, and are further processed to obtain highly accurate position, velocity, attitude and heading measurements of the carrier under dynamic environments. 
     Referring to FIG. 18, the coremicro inertial measurement unit of the present invention comprises an angular rate producer c 5  for producing three-axis (X axis, Y axis and Z axis) angular rate signals; an acceleration producer c 10  for producing three-axis (X-axis, Y axis and Z axis) acceleration signals; and an angular increment and velocity increment producer c 6  for converting the three-axis angular rate signals into digital angular increments and for converting the input three-axis acceleration signals into digital velocity increments. 
     Moreover, a position and attitude processor c 80  is adapted to further connect with the coremicro IMU of the present invention to compute position, attitude and heading angle measurements using the three-axis digital angular increments and three-axis velocity increments to provide a user with a rich motion measurement to meet diverse needs. 
     The position, attitude and heading processor c 80  further comprises two optional running modules: 
     (1) Attitude and Heeding, Module c 81 , producing attitude and heading angle only; and 
     (2) Position, Velocity, Attitude, and Heading Module c 82 , producing position, velocity, and attitude angles. 
     In general, the angular rate producer c 5  and the acceleration producer c 10  are very sensitive to a variety of temperature environments. In order to improve measurement accuracy, referring to FIG. 19, the present invention further comprises a thermal controlling means for maintaining a predetermined operating temperature of the angular rate producer c 5 , the acceleration producer c 10  and the angular increment and velocity increment producer c 6 . It is worth to mention that if the angular rate producer c 5 , the acceleration producer c 10  and the angular increment and velocity increment producer c 6  are operated in an environment under perfect and constant thermal control, the thermal controlling means can be omitted. 
     According to the preferred embodiment of the present invention, as shown in FIGS. 12, the thermal controlling means comprises a thermal sensing producer device c 15 , a heater device c 20  and a thermal processor c 30 . 
     The thermal sensing producer device c 15 , which produces temperature signals, is processed in parallel with the angular rate producer c 5  and the acceleration producer c 10  for maintaining a predetermined operating temperature of the angular rate producer c 5  and the acceleration producer c 10  and angular increment and velocity increment producer c 6  of the coremicro IMU, wherein the predetermined operating temperature is a constant designated temperature selected between 150° F. and 185° F., preferable 176° F. (±0.1° F.). 
     The temperature signals produced from the thermal sensing producer device c 15  are input to the thermal processor c 30  for computing temperature control commands using the temperature signals, a temperature scale factor, and a predetermined operating temperature of the angular rate producer c 5  and the acceleration producer c 10 , and produce driving signals to the heater device c 20  using the temperature control commands for controlling the heater device c 20  to provide adequate heat for maintaining the predetermined operating temperature in the coremicro IMU. 
     Temperature characteristic parameters of the angular rate producer c 5  and the acceleration producer c 10  can be determined during a series of the angular rate producer and acceleration producer temperature characteristic calibrations. 
     Referring to FIG. 20, when the above thermal processor c 30  and the heater device c 20  are not provided, in order to compensate the angular rate producer and acceleration producer measurement errors induced by a variety of temperature environments, the coremicro IMU of the present intention can alternatively comprise a temperature digitizer c 18  for receiving the temperature signals produced from the thermal sensing producer device c 15  and outputting a digital temperature value to the position, attitude, and heading processor c 80 . As shown in FIG. 29, the temperature digitizer c 18  can be embodied to comprise an analog/digital converter c 182 . 
     Moreover, the position, attitude, and heading processor c 80  is adapted for accessing temperature characteristic parameters of the angular rate producer and the acceleration producer using a current temperature of the angular rate producer and the acceleration producer from the temperature digitizer c 18 , and compensating the errors induced by thermal effects in the input digital angular and velocity increments and computing attitude and heading angle measurements using the three-axis digital angular increments and three-axis velocity increments in the attitude and heading processor c 80 . 
     In most applications, the output of the angular rate producer c 5  and the acceleration producer c 10  are analog voltage signals. The three-axis analog angular rate voltage signals produced from the angular producer c 5  are directly proportional to carrier angular rates, and the three-axis analog acceleration voltage signals produced from the acceleration producer c 10  are directly proportional to carrier accelerations. 
     When the outputting analog voltage signals of the angular rate producer c 5  and the acceleration producer c 90  are too weak for the angular increment and velocity increment producer c 6  to read, the angular increment and velocity increment producer c 6  may employ amplifying means c 660  and c 665  for amplifying the analog voltage signals input from the angular rate producer c 5  and the acceleration producer c 10  and suppress noise signals residing within the analog voltage signals input from the angular rate producer c 5  and the acceleration producer c 10 , as shown in FIGS. 15 and 16. 
     Referring to FIG. 21, the angular increment and velocity increment producer c 6  comprises an angular integrating means c 620 , an acceleration integrating means c 630 , a resetting means c 640 , and an angular increment and velocity increment measurement means c 650 . 
     The angular integrating means c 620  and the acceleration integrating means c 630  are adapted for respectively integrating the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals for a predetermined time interval to accumulate the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals as an uncompensated three-axis angular increment and an uncompensated three-axis velocity increment for the predetermined time interval to achieve accumulated angular increments and accumulated velocity increments. The integration is performed to remove noise signals that are non-directly proportional to the carrier angular rate and acceleration within the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals, to improve the signal-to-noise ratio, and to remove the high frequency signals in the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals. The signals are directly proportional to the carrier angular rate and acceleration within the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals. 
     The resetting means forms an angular reset voltage pulse and a velocity reset voltage pulse as an angular scale and a velocity scale which are input into the angular integrating means c 620  and the acceleration integrating means c 630  respectively. 
     The angular increment and velocity increment measurement means c 650  is adapted for measuring the voltage values of the three-axis accumulated angular increments and the three-axis accumulated velocity increments with the angular reset voltage pulse and the velocity reset voltage pulse respectively to acquire angular increment counts and velocity increment counts as a digital form of the angular increment and velocity increment measurements respectively. 
     In order to output real three-angular increment and velocity increment values as an optional output format to substitute the voltage values of the three-axis accumulated angular increments and velocity increments, the angular increment and velocity increment measurement means c 650  also scales the voltage values of the three-axis accumulated angular and velocity increments into real three-axis angular and velocity increment voltage values. 
     In the angular integrating means c 620  and the acceleration integrating means c 630 , the three-axis analog angular voltage signals and the three-axis analog acceleration voltage signals are each reset to accumulate From a zero value at an initial point of every predetermined time interval. 
     As shown in FIG. 23, in general, the resetting means c 640  can be an oscillator c 66 , so that the angular reset voltage pulse and the velocity reset voltage pulse are implemented by producing a timing pulse by the oscillator c 66 . In applications, the oscillator c 66  can be built with circuits, such as Application Specific Integrated Circuits (ASIC) chip and a printed circuit board. 
     As shown in FIG. 24, the angular increment and velocity increment measurement means c 650 , which is adapted for measuring the voltage values of the three-axis accumulated angular and velocity increments, is embodied as an analog/digital converter c 650 ′. In other words, the analog/digital converter c 650 ′ substantially digitizes the raw three-axis angular increment and velocity increment voltage values into digital three-axis angular increment and velocity increments. 
     Referring to FIGS. 17 and 21, the amplifying means c 660  and c 665  of the angular increment and velocity increment producer c 6  are embodied by an angular amplifier circuit c 61  and an acceleration amplifier circuit c 67  respectively to amplify the three-axis analog angular rate voltage signals and the three-axis analog acceleration voltage signals to form amplified three-axis analog angular rate signals and amplified three-axis analog acceleration signals respectively. 
     The angular integrating means c 620  and the acceleration integrating means c 630  of the angular increment and velocity increment producer c 6  ale respectively embodied as an angular integrator circuit c 62  and an acceleration integrator circuit c 68  for receiving the amplified three-axis analog angular rate signals and the amplified three-axis analog acceleration signals from the angular and acceleration amplifier circuits c 61 , c 67  which are integrated to form the accumulated angular increments and the accumulated velocity increments respectively. 
     The analog/digital converter c 650 ′ of the angular increment and velocity increment producer c 6  further includes an angular analog/digital converter c 63 , a velocity analog/digital converter c 69  and an input/output interface circuit c 65 . 
     The accumulated angular increments output from the angular integrator circuit c 62  and the accumulated velocity increments output from the acceleration integrator circuit are input into the angular analog/digital converter c 63  and the velocity analog/digital converter c 69  respectively. 
     The accumulated angular increments are digitized by the angular analog/digital converter c 63  by measuring the accumulated angular increments with the angular reset voltage pulse to form digital angular measurements of voltage in terms of the angular increment counts which are output to the input/output interface circuit c 65  to generate digital three-axis angular increment voltage values. 
     The accumulated velocity increments are digitized by the velocity analog/digital converter c 69  by measuring the accumulated velocity increments with the velocity reset voltage pulse to form digital velocity measurements of voltage in terms of the velocity increment counts which are output to the input/output interface circuit c 65  to generate digital three-axis velocity increment voltage values. 
     Referring to FIGS. 12 and 18, in order to achieve flexible adjustment of the thermal processor c 30  for the thermal sensing producer device c 15  with analog voltage output and the heater device c 20  with analog input, the thermal processor c 30  can be implemented in a digital feedback controlling loop as shown in FIG.  25 . 
     The thermal processor c 30 , as shown in FIG. 25, comprises an analog/digital converter c 304  connected to the thermal sensing producer device c 15 , a digital/analog converter c 303  connected to the heater device c 20 , and a temperature controller c 306  connected with both the analog/digital converter c 304  and the digital/analog converter c 303 . The analog/digital converter c 304  inputs the temperature voltage signals produced by the thermal sensing producer device c 15 , wherein the temperature voltage signals are sampled in the analog/digital converter c 304  to sampled temperature voltage signals which are further digitized to digital signals and output to the temperature controller c 306 . 
     The temperature controller c 306  computes digital temperature commands using the input digital signals from the analog/digital converter c 304 , a temperature sensor scale factor, and a pre-determined operating temperature of the angular rate producer and acceleration producer, wherein the digital temperature commands are fed back to the digital/analog converter c 303 . 
     The digital/analog converter c 303  converts the digital temperature commands input from the temperature controller c 306  into analog signals which are output to the heater device c 20  to provide adequate heat for maintaining the predetermined operating temperature of the coremicro IMU of the present invention 
     Moreover, as shown in FIG. 26, if the voltage signals produced by the thermal sensing producer device c 15  are too weak for the analog/digital converter c 304  to read, the thermal processor c 30  further comprises a first amplifier circuit c 301  between the thermal sensing producer device c 15  and the digital/analog converter c 303 , wherein the voltage signals from the thermal sensing producer device c 15  is first input into the first amplifier circuit c 301  for amplifying the signals and suppressing the noise residing in the voltage signals and improving the signal-to-noise ratio, wherein the amplified voltage signals are then output to the analog/digital converter c 304 . 
     The heater device c 20  requires a specific driving current signal. In this case referring to FIG. 27, the thermal processor c 30  can further comprise a second amplifier circuit  302  between the digital/analog converter c 303  and heater device c 20  for amplifying the input analog signals from the digital/analog converter c 303  for driving the heater device c 20 . 
     In other words, the digital temperature commands input from the temperature controller c 306  are converted in the digital/analog converter c 303  into analog signals which are then output to the amplifier circuit c 302 . 
     Referring to FIG. 28, an input/output interface circuit c 305  is required to connect the analog/digital converter c 304  and digital/analog converter c 303  with the temperature controller c 306 . In this case, as shown in FIG. 28, the voltage signals are sampled in the analog/digital converter c 304  to form sampled voltage signals that are digitized into digital signals. The digital signals are output to the input/output interface circuit c 305 . 
     As mentioned above, the temperature controller c 306  is adapted to compute the digital temperature commands using the input digital temperature voltage signals from the input/output interface circuit c 305 , the temperature sensor scale factor, and the pre-determined operating temperature of the angular rate producer and acceleration producer, wherein the digital temperature commands are fed back to the input/output interface circuit c 305 . Moreover, the digital/analog converter c 303  further converts the digital temperature commands input from the input/output interface circuit c 305  into analog signals which are output to the heater device c 20  to provide adequate heat for maintaining the predetermined operating temperature of the coremicro IMU. 
     Referring to FIG. 29, as mentioned above, the thermal processor c 30  and the heater device c 20  as disclosed in FIGS. 12,  18 ,  19 ,  20 , and  21  can alternatively be replaced by the analog/digital converter c 182  connected to the thermal sensing producer device c 15  to receive the analog voltage output from the thermal sensing producer device c 15 . If the voltage signals produced by the thermal sensing producer device c 15  are too weak for the analog/digital converter c 182  to read, referring to FIG. 30, an additional amplifier circuit c 181  can be connected between the thermal sensing producer device c 15  and the digital/analog converter c 182  for amplifying the analog voltage signals and suppressing the noise residing in the voltage signals and improving the voltage signal-to-noise ratio, wherein the amplified voltage signals are output to the analog/digital converter c 182  and sampled to form sampled voltage signals that are further digitized in the analog/digital converters c 182  to form digital signals connected to the attitude and heading processor c 80 . 
     Alternatively, an input/output interface circuit c 183  can be connected between the analog/digital converter c 182  and the attitude and heading processor c 80 . In this case, referring to FIG. 31, the input amplified voltage signals are sampled to form sampled voltage signals that are further digitized in the analog/digital converters to form digital signals connected to the input/output interface circuit c 183  before inputting into the attitude and heading processor c 80 . 
     Referring to FIG. 18, the digital three-axis angular increment voltage values or real values and three-axis digital velocity increment voltage values or real values are produced and outputted from the angular increment and velocity increment producer c 6 . 
     In order to adapt to digital three-axis angular increment voltage values and three-axis digital velocity increment voltage values from the angular increment and velocity increment producer c 6 , the attitude and heading module c 81 , as shown in FIG. 32, comprises a coning correction module c 81 , wherein digital three-axis angular increment voltage values from the input/output interface circuit c 65  of the angular increment and velocity increment producer c 6  and coarse angular rate bias obtained from an angular rate producer and acceleration producer calibration constants table at a high data rate (short interval) are input into the coning correction module c 811 , which computes coning effect errors by using the input digital three-axis angular increment voltage values and coarse angular rate bias, and outputs three-axis coning effect terms and three-axis angular increment voltage values at a reduced data rate (long interval), which are called three-axis long-interval angular increment voltage values. 
     The attitude and heading module c 81  further comprises an angular rate compensation module c 812  and an alignment rotation vector computation module c 815 . In the angular rate compensation module c 812 , the coning effect errors and three-axis long-interval angular increment voltage values from the coning correction module c 811  and angular rate device misalignment parameters, fine angular rate bias, angular rate device scale factor, and coning correction scale factor from the angular rate producer and acceleration producer calibration constants table are connected to the angular rate compensation module c 812  for compensating definite errors in the three-axis long-interval angular increment voltage values using the coning effect errors, angular rate device misalignment parameters, fine angular rate bias, and coning correction scale factor, and transforming the compensated three-axis long-interval angular increment voltage values to real three-axis long-interval angular increments using the angular rate device scale factor. Moreover, the real three-axis angular increments are output to the alignment rotation vector computation module c 815 . 
     The attitude and heading module c 81  further comprises an accelerometer compensation module c 813  and a level acceleration computation module c 814 , wherein the three-axis velocity increment voltage values from the angular increment and velocity increment producer c 6  and acceleration device misalignment, acceleration device bias, and acceleration device scale factor from the angular rate producer and acceleration producer calibration constants table are connected to the accelerometer compensation module c 813  for transforming the three-axis velocity increment voltage values into real three-axis velocity increments using the acceleration device scale factor, and compensating the definite errors in three-axis velocity increments using the acceleration device misalignment, accelerometer bias, wherein the compensated three-axis velocity increments are connected to the level acceleration computation module c 814 . 
     By using the compensated three-axis angular increments from the angular rate compensation module c 812 , an east damping rate increment from an east damping rate computation module c 8110 , a north damping rate increment from a north damping rate computation module c 819 , and vertical damping rate increment from a vertical damping rate computation module c 818 , a quaternion, which is a vector representing rotation angle of the carrier, is updated, and the updated quaternion is connected to a direction cosine matrix computation module c 816  for computing the direction cosine matrix, by using the updated quaternion. 
     The computed direction cosine matrix is connected to the level acceleration computation module c 814  and an attitude and heading angle extract module c 817  for extracting attitude and heading angle using the direction cosine matrix from the direction cosine matrix computation module c 816 . 
     The compensated three-axis velocity increments are connected to the level acceleration computation module c 814  for computing level velocity increments using the compensated three-axis velocity increments from the acceleration compensation module c 814  and the direction cosine matrix from the direction cosine matrix computation module c 816 . 
     The level velocity increments are connected to the east damping rate computation module c 8110  for computing east damping rate increments using the north velocity increment of the input level velocity increments from the level acceleration computation module c 814 . 
     The level velocity increments are connected to the north damping rate computation module c 819  for computing north damping rate increments using the east velocity increment of the level velocity increments from the level acceleration computation module c 814 . 
     The heading angle from the attitude and heading angle extract module c 817  and a measured heading angle from the external heading sensor c 90  are connected to the vertical damping rate computation module c 818  for computing vertical damping rate increments. 
     The east damping rate increments, north damping rate increments, and vertical damping rate are fed back to the alignment rotation vector computation module c 815  to damp the drift of errors of the attitude and heading angles. 
     Alternatively, in order to adapt real digital three-axis angular increment values and real three-axis digital velocity increment values from the angular increment and velocity increment producer c 6 , referring to FIG. 32, the real digital three-axis angular increment values from the angular increment and velocity increment producer c 6  and coarse angular rate bias obtained from an angular rate producer and acceleration producer calibration constants table at a high data rate (short interval) are connected to the coning correction module c 811  for computing coning effect errors in the coning correction module c 811  using the digital three-axis angular increment values and coarse angular rate bias and outputting three-axis coning effect terms and three-axis angular increment values at reduced data rate (long interval), which are called three-axis long-interval angular increment values, into the angular rate compensation module c 812 . 
     The coning effect errors and three-axis long-interval angular increment values from the coning correction module c 811  and angular rate device misalignment parameters and fine angular rate bias from the angular rate producer and acceleration producer calibration constants table are connected to the angular rate compensation module c 812  for compensating definite errors in the three-axis long-interval angular increment values using the coning effect errors, angular rate device misalignment parameters, fine angular rate bias, and coning, correction scale factor, and outputting the real three-axis angular increments to the alignment rotation vector computation module c 815 . 
     The three-axis velocity increment values from the angular increment and velocity increment producer c 6  and acceleration device misalignment, and acceleration device bias from the angular rate producer and acceleration producer calibration are connected into the accelerometer compensation module c 813  for compensating the definite errors in three-axis velocity increments using the acceleration device misalignment, and accelerometer bias; outputting the compensated three-axis velocity increments to the level acceleration computation module c 814 . 
     It is identical to the above mentioned processing that the following modules use the compensated three-axis angular increments from the angular rate compensation module c 812  and compensated three-axis velocity increments from the acceleration compensation module c 813  to produce attitude and heading angle. 
     Referring to FIG. 20,  31 , and  32 , which use the temperature compensation method by means of the temperature digitizer c 18 , in order to adapt to digital three-axis angular increment voltage value and three-axis digital velocity increment voltage values from the angular increment and velocity increment producer c 6 , the digital three-axis angular increment voltage values from the angular increment and velocity increment producer c 6  and coarse angular rate bias obtained from an angular rate producer and acceleration producer calibration constants table at a high data rate (short interval) are connected to the coning correction module c 811  for computing coning effect errors in the coning correction module c 811  using the digital three-axis angular increment voltage values and coarse angular rate bias, and outputting three-axis coning effect terms and three-axis angular increment voltage values at a reduced data rate (long interval), which are called three-axis long-interval angular increment voltage values, into the angular rate compensation module c 812 . 
     The coning effect errors and three-axis long-interval angular increment voltage values from the coning correction module c 811  and angular rate device misalignment parameters, fine angular rate bias, angular rate device scale factor, coning correction scale factor from the angular rate producer and acceleration producer calibration constants table, the digital temperature signals from input/output interface circuit c 183 , and temperature sensor scale factor are connected to the angular rate compensation module c 812  for computing current temperature of the angular rate producer, accessing angular rate producer temperature characteristic parameters using the current temperature of the angular rate producer, compensating definite errors in the three-axis long-interval angular increment voltage values using the coning effect errors, angular rate device misalignment parameters, fine angular rate bias, and coning correction scale factor, transforming the compensated three-axis long-interval angular increment voltage values to real three-axis long-interval angular increments, compensating temperature-induced errors in the real three-axis long-interval angular increments using the angular rate producer temperature characteristic parameters, and outputting the real three-axis angular increments to the alignment rotation vector computation module c 815 . 
     The three-axis velocity increment voltage values from the angular increment and velocity increment producer c 6  and acceleration device misalignment, acceleration bias, acceleration device scale factor from the angular rate producer and acceleration producer calibration constants table, the digital temperature signals from the input/output interface circuit c 183  of the temperature digitizer c 18 , and temperature sensor scale factor are connected to the acceleration compensation module c 813  for computing current temperature of the acceleration producer, accessing acceleration producer temperature characteristic parameters using the current temperature of the acceleration producer, transforming the three-axis velocity increment voltage values into real three-axis velocity increments using the acceleration device scale factor, compensating the definite errors in the three-axis velocity increments using the acceleration device misalignment and acceleration bias, compensating temperature-induced errors in the real three-axis velocity increments using the acceleration producer temperature characteristic parameters, and outputting the compensated three-axis velocity increments to the level acceleration computation module c 814 . 
     It is identical to the above mentioned processing that the following modules use the compensated three-axis angular increments from the angular rate compensation module c 812  and compensated three-axis velocity increments from the acceleration compensation module c 813  to produce the attitude and heading angles. 
     Alternatively, referring to FIGS. 13,  24 , and  25 , which use the temperature compensation method, in order to adapt real digital three-axis angular increment values and real three-axis digital velocity increment values from the angular increment and velocity increment producer c 6 , the attitude and heading module c 81  can be further modified to accept the digital three-axis angular increment values from the angular increment and velocity increment producer c 6  and coarse angular rate bias obtained from an angular rate producer and acceleration producer calibration constants table at a high data rate (short interval) into the coning correction module c 811  for computing coning effect errors in the coning correction module c 811  using the input digital three-axis angular increment values and coarse angular rate bias, and outputting three-axis coning effect data and three-axis angular increment data at a reduced data rate (long interval), which are called three-axis long-interval angular increment values, into the angular rate compensation module c 812 . 
     The coning effect errors and three-axis long-interval angular increment values from the coning correction module c 811  and angular rate device misalignment parameters and fine angular rate bias from the angular rate producer and acceleration producer calibration constants table, the digital temperature signals from the input/output interface circuit c 183  and temperature sensor scale factor are connected to the angular rate compensation module c 812  for computing current temperature of the angular rate producer, accessing angular rate producer temperature characteristic parameters using the current temperature of the angular rate producer, compensating definite errors in the three-axis long-interval angular increment values using the coning correction errors, angular rate device misalignment parameters, fine angular rate bias, and coning correction scale factor, compensating temperature-induced errors in the real three-axis long-interval angular increments using the angular rate producer temperature characteristic parameters, and outputting the real three-axis angular increments to an alignment rotation vector computation module c 815 . 
     The three-axis velocity increment values from the input/output interface circuit c 65  and acceleration device misalignment and acceleration bias from the angular rate producer and acceleration producer calibration constants table, the digital temperature signals from the input/output interface circuit c 183  and temperature sensor scale factor are input into the acceleration compensation module c 813  for computing current temperature of the acceleration producer, accessing the acceleration producer temperature characteristic parameters using the current temperature of the acceleration producer, compensating the definite errors in the three-axis velocity increments using the input acceleration device misalignment, acceleration bias, compensating temperature-induced errors in the real three-axis velocity increments using the acceleration producer temperature characteristic parameters, and outputting the compensated three-axis velocity increments to the level acceleration computation module c 814 . 
     It is identical to the above mentioned processing that the following modules use the compensated three-axis angular increments from the angular rate compensation module c 812  and compensated three-axis velocity increments from the acceleration compensation module c 813  to produce the attitude and heading angles. 
     Referring to FIG. 33, the Position, velocity, and attitude Module c 82  comprises: 
     a coning correction module c 8201 , which is same as the coning correction module c 811  of the attitude and heading module c 81 ; 
     an angular rate compensation module c 8202 , which is same as the angular rate compensation module c 812  of the attitude and heading module c 81 ; 
     an alignment rotation vector computation module c 8205 , which is same as the alignment rotation vector computation module c 815  of the attitude and heading module c 81 ; 
     a direction cosine matrix computation module c 8206 , which is same as the Direction cosine matrix computation module c 816  of the attitude and heading module c 81 ; 
     an acceleration compensation module c 8203 , which is same as the acceleration compensation module c 813  of the attitude and heading module c 81 ; 
     a level acceleration computation module c 8204 , which is same as the acceleration compensation module c 814  of the attitude and heading module c 81 ; and 
     an attitude and heading angle extract module c 8209 , which is same as the attitude and heading angle extract module c 817  of the attitude and heading module c 81 . 
     A position and velocity update module c 8208  accepts the level velocity increments from the level acceleration computation module c 8204  and computes position and velocity solution. 
     An earth and carrier rate computation module c 8207  accepts the position and velocity solution from the position and velocity update module c 8208  and computes the rotation rate vector of the local navigation frame (n frame) of the carrier relative to the inertial frame (i frame), which is connected to the alignment rotation vector computation module c 8205 . 
     In order to meet the diverse requirements of application systems, referring to FIGS. 21 and 24, the digital three-axis angular increment voltage values, the digital three-axis velocity increment, and digital temperature signals in the input/output interface circuit c 65  and the input/output interface circuit c 305  can be ordered with a specific format required by an external user system, such as RS-232 serial communication standard, RS-422 serial communication standard, the popular PCI/ISA bus standard, and 1553 bus standard, etc. 
     In order to meet diverse requirements of application systems, referring to FIGS. 13,  23  and  26 , the digital three-axis angular increment values, the digital three-axis velocity increment, and attitude and heading data in the input/output interface circuit c 85  are ordered with a specific format required by an external user system, such as RS-232 serial communication standard, RS-422 serial communication standard, PCI/ISA bus standard, and 1553 bus standard, etc. 
     As mentioned above, one of the key technologies of the present invention to achieve the coremicro IMU with a high degree of performance is to utilize a micro size angular rate producer, wherein the micro-size angular rate producer with MEMS technologies and associated mechanical supporting structure and circuitry board deployment of the coremicro IMU of the present invention are disclosed in the following description. 
     Another of the key technologies of the present invention to achieve the coremicro IMU with low power consumption is to design a micro size circuitry with small power consumption, wherein the conventional AISC (Application Specific Integrated Circuit) technologies can be utilized to shrink a complex circuitry into a silicon chip. 
     Existing MEMS technologies, which are employed into the micro size angular rate producer, use vibrating inertial elements (a micromachine) to sense vehicle angular rate via the Coriolis Effect. The angular rate sensing principle of Coriolis Effect is the inspiration behind the practical vibrating angular rate sensors. 
     The Coriolis Effect can be explained by saying that when an angular rate is applied to a translating or vibrating inertial element, a Coriolis force is generated. When this angular rate is applied to the axis of an oscillating inertial element, its tines receive a Coriolis force, which then produces torsional forces about the sensor axis. These forces are proportional to the applied angular rate, which then can be measured. 
     The force (or acceleration), Coriolis force (or Coriolis acceleration) or Coriolis effect, is originally named from a French physicist and mathematician, Gaspard de Coriolis (1792-1843), who postulated his acceleration in 1835 as a correction for the earth&#39;s rotation in ballistic trajectory calculations. The Coriolis acceleration acts on a body that is moving around a point with a fixed angular velocity and moving radially as well. 
     The basic equation defining Coriolis force is expressed as follows: 
     
       
           {right arrow over (F)}   Coriolis   =m{right arrow over (α)}   Coriolis =2 m ({right arrow over (ω)}×{right arrow over (V)} Oscillation ) 
       
     
     where 
     {right arrow over (F)} Coriolis  is the detected Coriolis force; 
     m is the mass of the inertial element; 
     {right arrow over (α)} Coriolis  is the generated Coriolis acceleration; 
     {right arrow over (ω)} is the applied (input) angular rotation rate; 
     {right arrow over (V)} Oscillation  is the oscillation velocity in a rotating frame. 
     The Coriolis force produced is proportional to the product of the mass of the inertial element, the input rotation rate, and the oscillation velocity of the inertial element that is perpendicular to the input rotation rate. 
     The major problems with micromachined vibrating type angular rate producer are insufficient accuracy, sensitivity, and stability. Unlike MEMS acceleration producers that are passive devices, micromachined vibrating type angular rate producer are active devices. Therefore, associated high performance electronics and control should be invented to effectively use hands-on micromachined vibrating type angular rate producers to achieve high performance angular rate measurements in order to meet the requirement of the coremicro IMU. 
     Therefore, in order to obtain angular rate signals signals from a vibrating type angular rate detecting unit, a dither drive signal or energy must be fed first into the vibrating type angular rate detecting unit to drive and maintain the oscillation of the inertial elements with a constant momentum. The performance of the dither drive signals is critical for the whole performance of a MEMS angular rate producer. 
     As shown in FIG.  34  and FIG. 35, which are a perspective view and a sectional view of the coremicro IMU of the present invention as shown in the block diagram of FIG. 18., the coremicro IMU comprises a first circuit board c 2 , a second circuit board c 4 , a third circuit board c 7 , and a control circuit board c 9  arranged inside a metal cubic case c 1 . 
     The first circuit board c 2  is connected with the third circuit board c 7  for producing X axis angular sensing signal and Y axis acceleration sensing signal to the control circuit board c 9 . 
     The second circuit board c 4  is connected with the third circuit board c 7  for producing Y axis angular sensing signal and X axis acceleration sensing signal to the control circuit board c 9 . 
     The third circuit board c 7  is connected with the control circuit board c 9  for producing Z axis angular sensing signal and Z axis acceleration sensing signals to the control circuit board c 9 . 
     The control circuit board c 9  is connected with the first circuit board c 2  and then the second circuit board c 4  through the third circuit board c 7  for processing the X axis, Y axis and Z axis angular sensing signals and the X axis, Y axis and Z axis acceleration sensing signals from the first, second and control circuit board to produce digital angular increments and velocity increments, position, velocity, and attitude solution. 
     As shown in FIG. 36, the angular producer c 5  of the preferred embodiment of the present invention comprises: 
     a X axis vibrating type angular rate detecting unit c 21  and a first front-end circuit c 23  connected on the first circuit board c 2 ; 
     a Y axis vibrating type angular rate detecting unit c 41  and a second front-end circuit c 43  connected on the second circuit board c 4 ; 
     a Z axis vibrating type angular rate detecting unit c 71  and a third front-end circuit c 73  connected on the third circuit board c 7 ; 
     three angular signal loop circuitries c 921 , which are provided in a ASIC chip c 92  connected on the control circuit board c 9 , for the first, second and third circuit boards c 2 , c 4 , c 7  respectively; 
     three dither motion control circuitries c 922 , which are provided in the ASIC chip c 92  connected on the control circuit board c 9 , for the first, second and third circuit boards c 2 , c 4 , c 7  respectively; 
     an oscillator c 925  adapted for providing reference pickoff signals for the X axis vibrating type angular rate detecting unit c 21 , the Y axis vibrating type angular rats detecting unit c 41 , the Z axis vibrating type angular rate detecting unit c 71 , the angle signal loop circuitry c 921 , and the dither motion control circuitry c 922 ; and 
     three dither motion processing modules c 912 , which run in a DSP (Digital Signal Processor) chipset c 91  connected on the control circuit board c 9 , for the first, second and third circuit boards c 2 , c 4 , c 7  respectively. 
     The first, second and third front-end circuits c 23 , c 43 , c 73 , each of which is structurally identical, are used to condition the output signal of the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  respectively and each further comprises: 
     a trans impedance amplifier circuit c 231 , c 431 , c 731 , which is connected to the respective X axis, Y axis or Z axis vibrating type angular rate detecting unit c 21 , c 41 , c 71  for changing the output impedance of the dither motion signals from a very high level, greater than 100 million ohms, to a low level, less than 100 ohms to achieve two dither displacement signals, which are A/C voltage signals representing the displacement between the inertial elements and the anchor combs. The two dither displacement signals are output to the dither motion control circuitry c 922 ; and 
     a high-pass filter circuit c 232 , c 432 , c 732 , which is connected with the respective X axis, Y axis or Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  for removing residual dither drive signals and noise from the dither displacement differential signal to form a filtered dither displacement differential signal to the angular signal loop circuitry c 921 . 
     Each of the X axis, Y axis and Z axis angular rate detecting units c 21 , c 41 , and c 71  is structurally identical except that sensing axis of each angular rate detecting unit is placed in an orthogonal direction. The X axis angular rate detecting unit c 21  is adapted to detect the angular rate of the vehicle along X axis. The Y axis angular rate detecting unit c 21  is adapted to detect the angular rate of the vehicle along Y axis. The Z axis angular rate detecting unit c 21  is adapted to detect the angular rate of the vehicle along Z axis. 
     Each of the X axis, Y axis and Z axis angular rate detecting units c 21 , c 41  and c 71  is a vibratory device, which comprises at least one set of vibrating inertial elements, including tuning forks, and associated supporting structures and means, including capacitive readout means, and uses Coriolis effects to detect vehicle angular rate. 
     Each of the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  receives signals as follows: 
     1) dither drive signals from the respective dither motion control circuitry c 922 , keeping the inertial elements oscillating; and 
     2) carrier reference oscillation signals from the oscillator c 925 , including capacitive pickoff excitation signals. 
     Each of the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  detects the angular motion in X axis, Y axis and Z axis respectively of a vehicle in accordance with the dynamic theory (Coriolis force), and outputs signals as follows: 
     1) angular motion-induced signals, including rate displacement signals which may be modulated carrier reference oscillation signals to a trans Impedance amplifier circuit c 231 , c 431 , c 731  of the first, second, and third front-end circuit c 23 ; and 
     2) its inertial element dither motion signals, including dither displacement signals, to the high-pass filter c 232 , c 432 , c 732  of the first, second, and third front-end circuit c 23 . 
     The three dither motion control circuitries c 922  receive the inertial element dither motion signals from the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  respectively, reference pickoff signals from the oscillator c 925 , and produce digital inertial element displacement signals with known phase. 
     In order to convert the inertial element dither motion signals from the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  to processible inertial element dither motion signals, referring to FIG. 41, each of the dither motion control circuitries c 922  comprises: 
     an amplifier and summer circuit c 9221  connected to the trans impedance amplifier circuit c 231 , c 431 , c 731  of the respective first, second or third front-end circuit c 23 , c 43 , c 73  for amplifying the two dither displacement signals for more than ten times and enhancing the sensitivity for combining the two dither displacement signals to achieve a dither displacement differential signal by subtracting a center anchor comb signal with a side anchor comb signal; 
     a high-pass filter circuit c 9222  connected to the amplifier and summer circuit c 9221  for removing residual dither drive signals and noise from the dither displacement differential signal to form a filtered dither displacement differential signal; 
     a demodulator circuit c 9223  connected to the high-pass filter circuit c 2225  for receiving the capacitive pickoff excitation signals as phase reference signals from the oscillator c 925  and the filtered dither displacement differential signal from the high-pass filter c 9222  and extracting the in-phase portion of the filtered dither displacement differential signal to produce an inertial element displacement signal with known phase; 
     a low-pass filter c 9225  connected to the demodulator circuit c 9223  for removing high frequency noise from the inertial element displacement signal input thereto to form a low frequency inertial element displacement signal; 
     an analog/digital converter c 9224  connected to the low-pass filter c 9225  for converting the low frequency inertial element displacement analog signal to produce a digitized low frequency inertial element displacement signal to the dither motion processing module c 912  (disclosed in the following text) running the DSP chipset c 91 ; 
     a digital/analog converter c 9226  processing the selected amplitude from the dither motion processing module c 912  to form a dither drive signal with the correct amplitude; and 
     an amplifier c 9227  which generates and amplifies the dither drive signal to the respective X axis, Y axis or Z axis vibrating type angular rate detecting unit c 21 , c 41 , c 71  based on the dither drive signal with the selected frequency and correct amplitude. 
     The oscillation of the inertial elements residing inside each of the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  is generally driven by a high frequency sinusoidal signal with precise amplitude. It is critical to provide the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 , c 41 , c 71  with high performance dither drive signals to achieve keen sensitivity and stability of X-axis, Y-axis and Z axis angular rate measurements. 
     The dither motion processing module c 912  receives digital inertial element displacement signals with known phase from the analog/digital converter c 9224  of the dither motion control circuitry c 922  for: 
     (1) finding the frequencies which have the highest Quality Factor (Q) Values, 
     (2) locking the frequency, and 
     (3) locking the amplitude to produce a dither drive signal, including high frequency sinusoidal signals with a precise amplitude, to the respective X axis, Y axis or Z axis vibrating type angular rate detecting unit c 21 , c 41 , c 71  to keep the inertial elements oscillating at the pre-determined resonant frequency. 
     The three dither motion processing modules c 912  is to search and lock the vibrating frequency and amplitude of the inertial elements of the respective X axis, Y axis or Z axis vibrating type angular rate detecting unit c 21 , c 41 , c 71 . Therefore, the digitized low frequency inertial element displacement signal is first represented in terms of its spectral content by using discrete Fast Fourier Transform (FFT). 
     Discrete Fast Fourier Transform (FFT) is an efficient algorithm for computing discrete Fourier transform (DFT), which dramatically reduces the computation load imposed by the DFT. The DFT is used to approximate the Fourier transform of a discrete signal. The Fourier transform, or spectrum, of a continuous signal is defined as: 
     
       
           X ( j ω)=∫ −∞   ∞   x ( t ) e   −jωt   dt   
       
     
     The DFT of N samples of a discrete signals X(nT) is given by:            X   s          (     k                 ω     )       =       ∑     n   =   0       N   -   1                         x        (   nT   )                   -   jω                   Tnk                           
     where ω=2π/NT, T is the inter-sample time interval. The basic property of FFT is its ability to distinguish waves of different frequencies that have been additively combined. 
     After the digitized low frequency inertial element displacement signals are represented in terms of their spectral content by using discrete Fast Fourier Transform (FFT), Q (Quality Factor) Analysis is applied to their spectral content to determine the frequency with global maximal Q value. The vibration of the inertial elements of the respective X axis, Y axis or Z axis vibrating type angular rate detecting unit c 21 , c 41 , c 71  at the frequency with global maximal Q value can result in minimal power consumption and cancel many of the terms that affect the excited mode. The Q value is a function of basic geometry, material properties, and ambient operating conditions. 
     A phase-locked loop and digital/analog converter is further used to control and stabilize the selected frequency and amplitude. 
     Referring to FIG. 43, the dither motion processing module c 912  further includes a discrete Fast Fourier Transform (FFT) module c 9121 , a memory array of frequency and amplitude data module c 9122 , a maxima detection logic module c 9123 , and a Q analysis and selection logic module c 9124  to find the frequencies which have the highest Quality Factor (Q) Values. 
     The discrete Fast Fourier Transform (FFT) module c 9121  is arranged for transforming the digitized low frequency inertial element displacement signal from the analog/digital converter c 9224  of the dither motion control circuitry c 922  to form amplitude data with the frequency spectrum of the input inertial element displacement signal. 
     The memory array of frequency and amplitude data module c 9122  receives the amplitude data with frequency spectrum to form an array of amplitude data with frequency spectrum. 
     The maxima detection logic module c 9123  is adapted for partitioning the frequency spectrum from the array of the amplitude data with frequency into plural spectrum segments, and choosing those frequencies with the largest amplitudes in the local segments of the frequency spectrum. 
     The Q analysis and selection logic module c 9124  is adapted for performing Q analysis on the chosen frequencies to select frequency and amplitude by computing the ratio of amplitude/bandwidth, wherein the range for computing bandwidth is between +−½ of the peek for each maximum frequency point. 
     Moreover, the dither motion processing module c 912  further includes a phase-lock loop c 9125  to reject noise of the selected frequency to form a dither drive signal with the selected frequency, which serves as a very narrow bandpass filter, locking the frequency. 
     The three angle signal loop circuitries c 921  receive the angular motion-induced signals from the X axis, Y axis and Z axis vibrating type angular rate detecting units c 21 . c 41 , c 71  respectively, reference pickoff signals from the oscillator c 925 , and transform the angular motion-induced signals into angular rate signals. Referring to FIG. 40, each of the angle signal loop circuitries c 921  for the respective first, second or third circuit board c 2 , c 4 , c 7  comprises: 
     a voltage amplifier circuit c 9211 , which amplifies the filtered angular motion-induced signals from the high-pass filter circuit c 232  of the respective first, second or third front-end circuit c 23 , c 43 , c 73  to an extent of at least 100 millivolts to form amplified angular motion-induced signals; 
     an amplifier and summer circuit c 9212 , which subtracts the difference between the angle rates of the amplified angular motion-induced signals to produce a differential angle rate signal; 
     a demodulator c 9213 , which is connected to the amplifier and summer circuit c 9212 , extracting the amplitude of the in-phase differential angle rate signal from the differential angle rate signal and the capacitive pickoff excitation signals from the oscillator c 925 ; 
     a low-pass filter c 9214 , which is connected to the demodulator c 9213 , removing the high frequency noise of the amplitude signal of the in-phase differential angle rate signal to form the angular rate signal output to the angular increment and velocity increment producer c 6 . 
     Referring to FIGS. 27 to  29 , the acceleration producer c 10  of the preferred embodiment of the present invention comprises: 
     a X axis accelerometer c 42 , which is provided on the second circuit board c 4  and connected with the angular increment and velocity increment producer  6  provided in the AISC chip c 92  of the control circuit board c 9 ; 
     a Y axis accelerometer c 22 , which is provided on the first circuit board c 2  and connected with angular increment and velocity increment producer c 6  provided in the AISC chip c 92  of the control circuit board c 9 ; and 
     a Z axis accelerometer c 72 , which is provided on the third circuit board  7  and connected with angular increment and velocity increment producer  6  provided in the AISC chip c 92  of the control circuit board c 9 . 
     Referring to FIG. 19,  35  and FIG. 36, thermal sensing producer device c 15  of the preferred embodiment of the present invention further comprises: 
     a first thermal sensing producing unit c 24  for sensing the temperature of the X axis angular rate detecting unit c 21  and the Y axis accelerometer c 22   
     a second thermal sensing producer c 44  for sensing the temperature of the Y axis angular rate detecting unit c 41  and the X axis accelerometer c 42 ; and 
     a third thermal sensing producer c 74  for sensing the temperature of the Z axis angular rate detecting unit c 71  and the Z, axis accelerometer c 72 . 
     Referring to FIG. 19 and 36, the heater device c 20  of the preferred embodiment of the present invention further comprises: 
     a first heater c 25 , which is connected to the X axis angular rate detecting unit c 21 , the Y axis accelerometer c 22 , and the first front-end circuit c 23 , for maintaining the predetermined operational temperature of the X axis angular rate detecting unit c 21 , the Y axis accelerometer c 22 , and the first front-end circuit c 23 ; 
     a second heater c 45 , which is connected to the Y axis angular rate detecting unit c 41 , the X axis accelerometer c 42 , and the second front-end circuit c 43 , for maintaining the predetermined operational temperature of the X axis angular rate detecting unit c 41 , the X axis accelerometer c 42 , and the second front-end circuit c 43 ; and 
     a third heater c 75 , which is connected to the Z axis angular rate detecting unit c 71 , the Z axis accelerometer c 72 , and the third front-end circuit c 73 , for maintaining the predetermined operational temperature of the Z axis angular rate detecting unit c 71 , the Z axis accelerometer c 72 , and the third front-end circuit c 73 . 
     Referred to FIGS. 12,  28 ,  29 ,  31 , and  35 , the thermal processor c 30  of the preferred embodiment of the present invention further comprises three identical thermal control circuitries c 923  and the thermal control computation modules c 911  running the DSP chipset c 91 . 
     As shown in FIGS. 29 and 35, each of the thermal control circuitries c 923  further comprises: 
     a first amplifier circuit c 9231 , which is connected with the respective X axis, a axis or Z axis thermal sensing producer c 24 , c 44 , c 74 , for amplifying the signals and suppressing the noise residing in the temperature voltage signals from the respective X axis, Y axis or Z axis thermal sensing producer c 24 , c 44 , c 74 &#39;and improving the signal-to-noise ratio; 
     an analog/digital converter c 9232 , which is connected with the amplifier circuit c 9231 , for sampling the temperature voltage signals and digitizing the sampled temperature voltage signals to digital signals, which are output to the thermal control computation module c 911 ; 
     a digital/analog converter c 9233  which converts the digital temperature commands input from the thermal control computation module c 911  into analog signals; and 
     a second amplifier circuit c 9234 , which receives the analog signals from the digital/analog converter  9233 , amplifying the input analog signals from the digital/analog converter c 9233  for driving the respective first, second or third heater c 25 , c 45 , c 75 ; and closing the temperature controlling loop. 
     The thermal control computation module c 911  computes digital temperature commands using the digital temperature voltage signals from the analog/digital converter c 9232 , the temperature sensor scale factor, and the pre-determined operating temperature of the angular rate producer and acceleration producer, wherein the digital temperature commands are connected to the digital/analog converter c 9233 . 
     In order to achieve a high degree of full functional performance for the coremicro IMU, a specific package of the first circuit board c 2 , the second circuit board c 4 , the third circuit board c 7 , and the control circuit board c 9  of the preferred embodiment of the present invention is provided and disclosed as follows: 
     In the preferred embodiment of the present invention, as shown in FIGS. 34,  30 , and  31 , the third circuit board c 7  is bonded to a supporting structure by means of a conductive epoxy, and the first circuit board c 2 , the second circuit board c 4 , and the control circuit board c 9  are arranged in parallel to bond to the third circuit board c 7  perpendicularly by a non conductive epoxy. 
     In other words, the first circuit board c 2 , the second circuit board c 4 , and the control circuit board c 9  are soldered to the third circuit board c 7  in such a way as to use the third circuit board c 7  as an interconnect board, thereby avoiding the necessity to provide interconnect wiring, so as to minimize the small size. 
     The first, second, third, and control circuit boards c 2 , c 4 , c 7 , and c 9  are constructed using ground planes which are brought out to the perimeter of each circuit board c 2 , c 4 , c 7 , c 9 , so that the conductive epoxy can form a continuous ground plane with the supporting structure. In this way the electrical noise levels are minimized and the thermal gradients are reduced. Moreover, the bonding process also reduces the change in misalignments due to structural bending caused by acceleration of the IMU.