Patent Publication Number: US-2019173393-A1

Title: Voltage balancing of voltage source converters

Description:
BACKGROUND OF THE DISCLOSURE 
     This application relates to a voltage source converter and to methods and apparatus for control of a director switch of a voltage source converter for voltage balancing, and especially to a voltage source converter for use in high voltage power distribution and in particular to a voltage source converter having converter arms with a director switch having multiple switching elements each having an associated clamp capacitor. 
     HVDC (high-voltage direct current) electrical power transmission uses direct current for the transmission of electrical power. This is an alternative to alternating current electrical power transmission which is more common. There are a number of benefits to using HVDC electrical power transmission. 
     In order to use HVDC electrical power transmission, it is typically necessary to convert alternating current (AC) to direct current (DC) and back again. To date most HVDC transmission systems have been based on line commutated converters (LCCs), for example such as a six-pulse bridge converter using thyristor valves. LCCs use elements such as thyristors that can be turned on by appropriate trigger signals and remain conducting as long as they are forward biased and current remains flowing. In LCCs the converter relies on the connected AC voltage to provide commutation from one valve to another. 
     Increasingly however voltage source converters (VSCs) are being proposed for use in HVDC transmission. HVDCs use switching elements such as insulated-gate bipolar transistors (IGBTs) that can be controllably turned on and turned off independently of any connected AC system. VSCs are thus sometimes referred to as self-commutating converters. 
     VSCs typically comprise multiple converter arms, each of which connects one DC terminal to one AC terminal. For a typical three phase AC input/output there are six converter arms, with the two arms connecting a given AC terminal to the high and low DC terminals respectively forming a phase limb. Each converter arm comprises an apparatus which is commonly termed a valve and which typically comprises a plurality of elements which may be switched in a desired sequence. 
     In one form of known VSC, often referred to as a six switch converter, each valve comprises a set of series connected switching elements, typically insulated gate bipolar transistors (IGBTs) connected with respective antiparallel diodes. The IGBTs of the valve are switched together to electrically connect or disconnect the relevant AC and DC terminals, with the valves of a given phase limb typically being switched in anti-phase. By using a pulse width modulated (PWM) type switching scheme for each arm, conversion between AC and DC voltage can be achieved. 
     In another known type of VSC, referred to a modular multilevel converter (MMC), each valve comprises a chain-link circuit having a plurality of cells connected in series, each cell comprising an energy storage element such as a capacitor and a switch arrangement that can be controlled so as to either connect the energy storage element between the terminals of the cell or bypass the energy storage element. The cells are sometimes referred to as sub-modules, with a plurality of cells forming a module. The sub-modules of a valve are controlled to connect or bypass their respective energy storage elements at different times so as to vary over the time the voltage difference across the plurality of cells. By using a relatively large number of sub-modules and timing the switching appropriately the valve can synthesise a stepped waveform that approximates to a desired waveform, such as a sine wave, to convert from DC to AC or vice versa with low levels of harmonic distortion. As the various sub-modules are switched individually and the changes in voltage from switching an individual sub-module are relatively small a number of the problems associated with the six switch converter are avoided. 
     In the MMC design each valve is operated continually through the AC cycle with the two valves of a phase limb being switched in synchronism to provide the desired voltage waveform. 
     Recently a variant converter has been proposed wherein a chain-link of a series of connected cells is provided in a converter arm for providing a stepped voltage waveform as described, but each converter arm is turned off for at least part of the AC cycle. Thus the plurality of series connected cells for voltage wave-shaping are connected in series with an arm switch, referred to as a director switch, which can be turned off when the relevant converter arm is in the off state and not conducting. Such a converter has been referred to as an Alternate-Arm-Converter (AAC). An example of such a converter is described in WO2010/149200. 
       FIG. 1  illustrates a known Alternate-Arm-Converter (AAC)  100 . The example converter  100  has three phase limbs  101   a - c , each phase limb having a high side converter arm connecting the relevant AC terminal  102   a - c  to the high side DC terminal DC+ and a low side converter arm connecting the relevant AC terminal  102   a - c  to the low side DC terminal DC−. Each converter arm comprises a circuit arrangement  103  of series connected cells, the arrangement  103  being in series with an arm switch  104  and inductances  105 . It will be noted that  FIG. 1  illustrates a single arm inductance but one skilled in the art will appreciated that the arm inductance may in practice be distributed along the arm between the AC and DC terminals. 
     The circuit arrangement  103  comprises a plurality of cells  106  connected in series. Each cell  106  has an energy storage element that can be selectively connected in series between the terminals of the cell or bypassed. In the example shown in  FIG. 1  each cell  106  has terminals  107   a,    107   b  for high-side and low-side connections respectively and comprises a capacitor  108  as an energy storage element. The capacitor  108  is connected with cell switching elements  109 , e.g. IGBTs with antiparallel diodes, to allow the terminals  107   a  and  107   b  of the cell to be connected via a path that bypasses capacitor  108  or via a path that includes capacitor  108  connected in series. In the example illustrated in  FIG. 1  each cell comprises four cell switching elements  109  in a full H-bridge arrangement such that the capacitor can be connected in use to provide either a positive or a negative voltage difference between the terminals  107   a  and  107   b.  In some embodiments however at least some of the cells may comprise switching elements in a half bridge arrangement such that the capacitor can be bypassed or connected to provide a voltage difference of a given polarity. The circuit arrangement  103  of such series connected cells can thus operate to provide a voltage level that can be varied over time to provide stepped voltage waveform for wave-shaping as discussed above. The circuit arrangement  103  is sometimes referred to as a chain-link circuit or chain-link converter or simply as a chain-link. In this disclosure the circuit arrangement  103  of such series connected cells for providing a controlled voltage shall be referred to as a chain-link. 
     In the AAC converter the chain-link  103  in each converter arm is connected in series with an arm switch  104 , which will be referred to herein as a director switch, which may comprise a plurality of series connected arm switching elements  110 . The director switch of a converter arm may for example comprise high voltage elements with turn-off capability such as IGBTs or the like with antiparallel diodes. When a particular converter arm is conducting, the chain-link  103  is switched in sequence to provide a desired waveform in a similar fashion as described above with respect to the MMC type converter. However in the AAC converter each of the converter arms of a phase limb is switched off for part of the AC cycle and during such a period the director switch  104  is turned off. 
     When the converter arm is thus in an off state and not conducting the voltage across the arm is shared between the director switch and the chain-link circuit. Compared to the MMC type VSC the required voltage range for the chain-link of each converter arm of an AAC type converter is thus reduced, with consequent savings in the cost and size of the converter. 
     As mentioned above the director switch  104  is typically formed from a plurality of series connected IGBTs. The IGBTs are typically connected in parallel with a balancing resistor for static voltage sharing between the individual switching elements of the director switch. In addition there may be a clamp snubber circuit located next to the IGBT to mitigate voltage overshoot during a turn-off transient event which comprises a capacitor and diode. 
     One issue that can arise in such an arrangement is a voltage imbalance across the switch elements of the director switch, e.g. the IGBTs and the associated clamp capacitors. 
     BRIEF SUMMARY 
     The present disclosure thus relates to methods and apparatus for control of voltage source converters that address issues of voltage imbalance. 
     Thus according to the present invention there is provided a method of controlling a director switch unit of a voltage source converter comprising a semiconductor switching element and an associated clamp capacitor connected across the semiconductor switching element, the method comprising, in a voltage balancing mode: operating the director switch unit in a voltage balancing mode such that power drawn from the clamp capacitor varies based on the voltage across the clamp capacitor. 
     By operating in a voltage balancing mode such that the power drawn from the clamp capacitor varies with the voltage of the clamp capacitor, rather than having a constant power characteristic, voltage balancing of the clamp capacitors of multiple serially connected director switch units can be achieved. In the voltage balancing mode the power demand for power drawn from the clamp capacitor may have the characteristic of a resistive load. Thus a greater current may be drawn as the voltage of the clamp capacitor increases ensuring that the clamp capacitors with the greatest voltages are discharged more than those with lower voltages. 
     In some examples the method involves, in the voltage balancing mode, varying the power demand of a power supply that draws power from the clamp capacitor, for instance a floating power supply, based on the voltage of the clamp capacitor. The voltage of the clamp capacitor may be compared to at least one voltage threshold. A power demand may be determined based on said comparison. In some examples the power demand may have a first value if the clamp capacitor voltage is below a first threshold and a second, higher, value if the clamp capacitor voltage is above the first threshold. 
     In some examples the method involves, in the voltage balancing mode, varying the current demand from the clamp capacitor based on the voltage of the clamp capacitor. In some examples the current demand may have a component proportional to the voltage of the clamp capacitor. The current demand may include a component which is inversely proportional to a virtual resistance value. The virtual resistance value may be selected to give a predetermined rated power at a maximum expected clamp capacitor voltage. In some examples the current demand may further have a component that varies based on an indication of the rate of change of the clamp capacitor voltage. The current demand may be controlled by controlling a controllable current source. 
     In some examples the method involves, in the voltage balancing mode, varying, controlling a crowbar circuit connected in parallel with the clamp capacitor so as to provide a predetermined average resistance value, wherein the crowbar circuit comprises a crowbar resistor in series with a semiconductor switching element. The semiconductor switching element of the crowbar circuit may be controlled in a duty cycle to provide the predetermined average resistance value. 
     Any additional power drawn which is in excess of the rated power demand of a power supply of the director switch unit may be stored in an energy storage reservoir of the power supply. In some instances the power supply may further comprise a balancing reservoir for storing power drawn in the voltage balancing mode. The method may comprise monitoring the voltage of the balancing reservoir and discharging the balancing reservoir when above a predetermined voltage level. 
     In some examples the director switch unit may comprise at least one loading resistor, the loading resistor having a resistance value such that, in use at a clamp capacitor voltage within the nominal operating range of the director switch unit, the current drawn from the clamp capacitor increases with clamp capacitor voltage. The at least one loading resistor may comprise a loading resistor in parallel with a snubber diode and/or a loading resistor in parallel with the clamp capacitor. 
     Also provided is a director switch unit of a voltage source converter comprising: a semiconductor switching element; and a clamp capacitor connected across the semiconductor switching element, the director switch unit being operable in a voltage balancing mode such that the power drawn from the clamp capacitor varies based on the voltage across the clamp capacitor. 
     Any of the variants of the method described above may be applicable to the director switch unit. In particular, in the voltage balancing mode, the director switch unit is configured such that the power demand for power drawn from the clamp capacitor has the characteristic of a resistive load. 
     The director switch unit may comprise a power supply, such as a floating power supply, that draws power from the clamp capacitor and a controller for controlling the power drawn from the clamp capacitor. 
     In some examples the controller may be configured to, in the voltage balancing mode, vary the power demand of the power supply based on the voltage of the clamp capacitor. The controller may be configured to compare the clamp capacitor voltage to at least one voltage threshold and determine a power demand based on said comparison. The controller may control the power demand to a first value if the clamp capacitor voltage is below a first threshold and to a second, higher, value if the clamp capacitor voltage is above the first threshold. 
     In some examples the controller may be configured to, in the voltage balancing mode, vary the current demand from the clamp capacitor based on the voltage of the clamp capacitor. The controller may be configured to control the current demand to have a component proportional to the voltage of the clamp capacitor. The current demand may have a component inversely proportional to a virtual resistance value. The virtual resistance value may be selected to give a predetermined rated power at a maximum expected clamp capacitor voltage. The current demand may further have a component that varies based on an indication of the rate of change of the clamp capacitor voltage. The director switch unit may include a controllable current source. The controller may be configured to vary the current demand by controlling the controllable current source. 
     In some examples the director switch unit comprises a crowbar circuit connected in parallel with the clamp capacitor and a controller for controlling the crowbar circuit so as to provide a predetermined average resistance value, wherein the crowbar circuit comprises a crowbar resistor in series with a semiconductor switching element. The controller may be configured to control the semiconductor switching element of the crowbar circuit in a duty cycle to provide the predetermined average resistance value. 
     The power supply may comprise an energy storage reservoir and the director switch unit may be configured such that any additional power drawn which is in excess of the rated power demand of the power supply of the director switch unit is stored in the energy storage reservoir. The power supply may further comprise a balancing reservoir for storing power drawn in the voltage balancing mode. A monitor may be provided for monitoring the voltage of the balancing reservoir and discharging the balancing reservoir when above a predetermined voltage level. 
     In some examples the director switch unit may comprise at least one loading resistor, the loading resistor having a resistance value such that, in use at a clamp capacitor voltage within the nominal operating range of the director switch unit, the current drawn from the clamp capacitor increases with clamp capacitor voltage. The at least one loading resistor may comprise a loading resistor in parallel with a snubber diode. The at least one loading resistor may comprise a loading resistor connected across or in parallel with the clamp capacitor. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention will now be described by way of example only with respect to the accompanying drawings, of which: 
         FIG. 1  illustrates one example of an alternate-arm-converter (AAC) type voltage source converter (VSC); 
         FIG. 2  illustrates one example of circuitry associated with a switching element of a director switch of a VSC; 
         FIG. 3  illustrates simulated results for a director switch operating with an initial voltage imbalance; 
         FIG. 4  illustrates the principles of varying the rate power of the floating power supply based on the clamp capacitor voltage; 
         FIG. 5  illustrates simulated results for a director switch operating according to strategy discussed with respect  FIG. 4 ; 
         FIG. 6  illustrates control over charging of the long term storage based on the change in clamp capacitor voltage; 
         FIG. 7  illustrates simulated results for a director switch operating according to another embodiment; 
         FIG. 8  illustrates PWM control of the crowbar circuit; 
         FIG. 9  illustrates simulated results for a director switch operating according to strategy discussed with respect to  FIG. 8 ; 
         FIG. 10  illustrates a director switch unit with an additional balancing reservoir; 
         FIG. 11  illustrates a converter arm according to an embodiment; 
         FIG. 12  illustrates current based control of the DC/DC converter of the floating power supply; and 
         FIGS. 13A and 13B  illustrates another example using loading resistors. 
     
    
    
     DETAILED DESCRIPTION 
     As mentioned above some types of voltage source converter (VSC), such as the alternate-arm-converter (AAC) illustrated in  FIG. 1 , comprise an arm switch  104  in a converter arm of the VSC for switching the converter arm between being conductive or non-conductive during the power cycle. Such an arm switch is referred to herein as a director switch. To provide the necessary voltage rating the director switch will typically comprise a plurality of switching elements  110  such as IGBTs connected in series. 
     In use the director switch of a converter arm may be turned off for part of the power cycle. In the AAC type converter, where each converter arm comprises a director switch  104  in series with a chain-link circuit  103  for voltage wave-shaping, the director switch of a converter arm is turned off at a desired point in the power cycle. In normal operation the director switch is turned off (or on) at a point when the chain-links of the converter arms of the phase limb are providing suitable voltages such that the voltage across the director switch being turned off (or on) is substantially zero. The two converter arms of a phase limb are also controlled together so that in normal operation there is substantially no current through the director switch at the point at which it is turned-off or on. 
     Once turned off the voltage across the director switch typically increases as the voltage at the AC terminal of the phase limb varies. To ensure correct voltage sharing between the various switching elements of the director switch there is typically a balancing resistor connected in parallel with each switch element. Additionally there is typically a clamp snubber circuit comprising at least a capacitor and a diode to mitigate voltage overshoot during a turn-off transient event. 
       FIG. 2  illustrates one example of a director switch unit  200  associated with an individual switching element  110  of a director switch. As used herein the term director switch unit shall be used to refer to an individual switching element  110  and its associated circuitry. The switching element  110  comprises a semiconductor switching element such as an IGBT with antiparallel diode. In parallel with the IGBT  110  is typically a balancing resistor  201 . A clamp snubber circuit comprises clamp capacitor  202  connected in parallel with the IGBT  110  via snubber diode  203 . In some instances a resistor (not shown) may be connected across the snubber diode  203  to provide for slow or long-term discharge of the capacitor voltage for safe handling, after power off. The IGBT  110  is controlled by local control electronics  204  including a suitable gate driver. There may also be a crowbar circuit in parallel with the switching element  110 , the crowbar comprising a parallel combination of resistor  205  and diode  206  in series with a semiconductor switch element such as transistor  207 . 
     It will be appreciated that  FIG. 2  only illustrates those components that are of interest for the present disclosure and in practice there may be may additional components and circuitry associated with the individual switching elements of the director switch such as crowbar circuits or surge arrestors for example. 
     Recently it has been proposed that the local control electronics may be powered by a floating power supply  208  that draws power from the capacitor  202  that forms part of the clamp snubber circuit. In the example illustrated in  FIG. 2  the floating power supply  208  thus comprises a first DC-DC converter  209  for converting from the high voltage across the IGBT/clamp capacitor  202  to an intermediate voltage and a second DC-DC converter  210  for converting to a low voltage suitable for powering the control electronics  204 , e.g. gate driver, for the IGBT  110 . The floating power supply may also comprise some long term energy storage  211  such as a suitable capacitor arrangement for providing power during fault events where there may temporarily be no voltage across the director switch. 
     The floating power supply  208  typically exhibits a constant power load characteristic, due to the control loop of the DC-DC converters  209  and  210  which keep the output voltage constant regardless of any variation in input voltage. Therefore, if the load current demanded by the gate driver of the control electronics  204  remains constant, as it is typically the case, the input current demanded by the floating power supply  208  will change to keep a constant product of voltage V and current I, i.e. a constant V*I product. Thus if the input voltage to the floating power supply, i.e. the voltage across the clamp capacitor  202 , increases or decreases, the input current will proportionally decrease or increase respectively. 
     It has been appreciated that this effect can contribute to any imbalance in voltage across the IGBTs and corresponding clamp capacitors of the director switch units of the director switch. If the voltage across a given IGBT  110  or its associated clamp capacitor  202  decreases then a greater current will be demanded by the floating power supply, which will contribute to a further voltage decrease. Conversely if the voltage across the IGBT/clamp capacitor is increasing then the input current demanded by the floating power supply will decrease, thus contributing to a further increase in the voltage across the IGBT/clamp capacitor. 
     Due to the relatively small capacitance of the clamp capacitor  202 , which may for instance be of the order of 1 μF, this effect may be significant. It has been found that for a director switch formed from director switch units which include a floating power supply arranged to take power from the clamp capacitor, then any component mismatch between the director switch units, e.g. between the voltage balancing resistors associated with different IGBTs, or any difference in the instantaneous voltage across the director switch units can lead to a relatively large voltage imbalance being generated over a number of cycles of VSC operation. 
     This is especially the case if the floating supply is  208  charging the long term storage  211 , when the power drawn by the floating supply is comparatively greater than that drawn during normal operation. 
       FIG. 3  illustrates waveforms for a simulated director switch having a voltage imbalance between the switching elements. The director switch was simulated with just first and second director switching units, with the first switching unit having a higher starting instantaneous voltage than the second switching unit, and the simulation covered several power cycles of the converter arm. The starting voltage of the clamp capacitor of the first switching unit was simulated at 1500V with the capacitor of the second switching unit at 500V. The floating power supply was simulated with a power draw of 22 W. 
     The top plot of  FIG. 3  illustrates the voltage waveform  301  across the whole director switch and the voltage waveforms  302  and  303  across the switching elements of the first and second switching units respectively. The middle plot shows the voltages  304  and  305  of the clamp capacitors of the first and second switching units respectively. The lower plot illustrates the current demand  306  of the floating power supply of the first switching unit and the current demand  307  of the floating power supply of the second switching unit. 
     It can be seen that the initial starting voltage difference quickly leads to an increasing voltage imbalance and the system quickly diverges to a situation of a high voltage imbalance. Also, the floating power supply of the first director switch unit, corresponding to the clamp capacitor with the highest voltage, takes less current than that of the second director switch unit with a lower voltage across it. 
     In the simulation results illustrated in  FIG. 2  the current demand in the floating supply corresponding to the clamp capacitor with less voltage across is repeatedly interrupted during a substantial part of the cycle. In this example the DC-DC converter stops operation if the input voltage drops below a threshold value, for instance to prevent the input current becoming too high at low input voltages or due to other deign considerations. 
     The effect of the floating power supply can thus be seen to exacerbate any initial voltage difference. This could finally result that an excess voltage is developed across some director switch units whilst there is a loss of power for the control electronics of others. 
     It will be well understood by one skilled in the art that the voltage of a clamp capacitor reaching high levels is potentially dangerous as it can cause the destruction of devices and circuit components. The director switching unit thus typically includes one or more protective measures such as the crowbar circuit. In the event of an over-voltage the transistor  207  of the crowbar circuit may be turned on to discharge the clamp capacitor via crowbar resistor  205 . There may additionally or alternatively be a surge protection device (not shown in  FIG. 2 ) in parallel with the switching element  110 . In some VSCs there may be an active clamping circuit built in the gate driver of each IGBT. 
     There may be thus some mechanism for coping with over-voltage of a clamp capacitor, however were a clamp capacitor voltage to fall below the operating threshold of the floating supply, this would cause the main DC-DC converter  209  to stop working. If during that time there is not enough energy stored in the long-term storage  211  to repeatedly maintain the operation of the gate driver supply, some IGBTs will be unexpectedly turned off, with a consequent risk of destruction. Since this imbalance situation may well arise when the long-term storage stage is being charged, it is very likely that the long-term storage capacitors will not have enough energy accumulated to support this situation. 
     Embodiments of the present invention thus relate to methods and apparatus for voltage balancing that at least mitigate some of the above mentioned issues and which can be used satisfactorily with switching units having a floating power supply that draws power from the clamp capacitor. As will be described in more detail below embodiments of the present invention may operate such that the load on the clamp capacitor, which comprises the floating power supply, behaves as a resistive load. 
     In a method according an embodiment of the present invention the director switching unit of a director switch is controlled such that the power drawn from the clamp capacitor, at least in a voltage balancing mode, varies based on the voltage across the clamp capacitor. The director switch may be operated such the power demand for power drawn from the clamp capacitor has the characteristic of a resistive load, rather than a constant power load. 
     As will be understood by one skilled in the art the properties of a resistive load are such that if the voltage across the load is increased, the current drawn by the load is also increased. In such a case were there to be a voltage imbalance between two clamp capacitors of two director switching units then a greater current would be drawn from the clamp capacitor with the highest voltage resulting in that capacitor discharging more rapidly than the other. The result would be to improve the voltage balance between the clamp capacitors, rather than exacerbate the difference was described above where a contrast power load is applied to the clamp capacitor. 
     There are various ways in which the power draw from the clamp capacitor can be controlled to provide the general properties of the resistive load. 
     In one embodiment the power demand from the floating power supply is varied based on an indication of the voltage level of the clamp capacitor. In one embodiment the voltage of the clamp capacitor may be determined and compared to at least one threshold level, with the rated power drawn by the floating power supply varies depending on whether the voltage of the clamp capacitor is above or below the threshold(s). 
       FIG. 4  illustrates the principle of altering the rated power of the floating power supply in this way.  FIG. 4  illustrates the instantaneous value of the voltage V CL  of a clamp capacitor of a director switching unit and how it varies over the course of the power cycle. This voltage level is compared to a voltage threshold  402 . At time when the voltage V CL  is below the threshold the floating power supply is configured to have a first rated power demand P 1 . If however the voltage V CL  exceeds the threshold the power rating is increased to a higher rated power P 2  so that the floating power supply draws more power. In this example the clamp capacitor voltage V CL  exceed threshold for part of the power cycle and thus the rated power demand of the floating power supply varies during the power cycle. 
     It will be appreciated that the operation of the floating power supply at any given power rating P 1  or P 2  is as a constant power load. However, because of the change of power rating during the cycle, over the course of a whole cycle the average power demand will follow resistive load profile. Clamp capacitors with a higher voltage level will, over the course of a cycle, draw a higher average power than clamp capacitors with a lower voltage level. In this case, the rated power level P 1  may correspond to the usual load of the gate driver and FPS electronics, whereas the higher power demand P 2  may be used to charge the Long Term Storage (LTS) capacitor stage, seen in  FIG. 2  as  211 . It is this change of power rating during the cycle that provides the voltage balancing mode. The power rating of the floating power supply be varied by a controller  212  that changes parameters of the floating power supply based on an indication of the voltage of the clamp capacitor. 
     This control strategy was applied for a simulated director switch as discussed above in relation to  FIG. 2 , i.e. a director switch with first and second director switching units, with the first switching unit having a higher starting instantaneous voltage than the second switching unit. Again the simulation covered several power cycles of the converter arm. The same starting voltage imbalance as  FIG. 2  was simulated, i.e. with the starting voltage of the clamp capacitor of the first switching unit at 1500V with the capacitor of the second switching unit at 500V. In this simulation however the floating power supply was simulated with a power draw that varied between 22 W and 2.2 W depending on whether the clamp capacitor voltage was above or below a voltage threshold of 800V. The results are illustrated in  FIG. 5 . 
     Again the top plot of  FIG. 5  shows the total voltage  501  across the director switch and the voltages across the first  502  and second  503  director switch units respectively. The middle plot shows the voltages of the first  504  and second  505  clamp capacitors and the lower plot shows the current draw of the floating power supplies of the first  506  and second  507  director switch units. It can be seen that the average current demand of the floating power supply of the first director switching unit is greater over the first power cycle than that off the second director switch unit. This reduces the voltage imbalance of the clamp capacitors and over the course of just two power cycles the voltage imbalance is effectively removed. 
     Note that as an alternative to changing the power demand level the current level could instead by varied based on the clamp capacitor voltage being compared to one or more threshold levels. It will be appreciated that there may be more than one threshold and more than two different power or current levels. 
     Additionally or alternatively the power demand of the floating power supply may be varied during the course of a power supply by only charging the long term storage  211  of the floating power supply  208  at a time when the clamp capacitor is effectively itself being charged by the voltage across the relevant director switch unit and thus there is plenty of energy available to sustain charging of the long term storage. 
     This is illustrated in  FIG. 6 .  FIG. 6  illustrates how the voltage waveform V DS  across a director switching unit of the director switch may evolve over time during the period over which the director switch of a converter arm is turned off. For example in an AAC type converter the director switch of the high-side converter arm of a phase limb may be turned off at or before the negative part of the relevant phase cycle. During the period the director switch is off the voltage across the director switch will typically increase to a maximum and then decrease back to zero at which point the director switch is turned on again. The voltage across the director switch will be shared across the director switching units in the off state and across the clamp capacitors. If the voltage across the director switching unit V DS  becomes higher than the present voltage of the clamp capacitor, the relevant clamp capacitor will start to be charged. During this period  602  the voltage change of the clamp capacitor, i.e. dV CL /dt, will be positive and there will be plenty of energy available for the floating power supply  208  to charge its long term storage without depleting the charge of the clamp capacitor. In some embodiments therefore the floating power supply may be configured so as to charge the long term storage of the floating power supply predominantly at times when the voltage of the clamp capacitor is decreasing. During periods when the voltage of the clamp capacitor is decreasing, i.e. dV CL /dt is negative, then charging of the long term storage of the floating power supply may be suspended. 
     In another embodiment a resistive load type profile is generated by drawing a current from the clamp capacitor with a component that varies with the voltage of the clamp capacitor. The current may in some embodiments be proportional to the voltage of the clamp capacitor. The current may be drawn inversely proportional to an emulated resistance value, for example a controllable current source may be configured to draw a current that varies in accordance with the clamp capacitor voltage and a virtual resistor value. This value of this virtual resistor R VIRTUAL  could for instance be chosen depending on the maximum expected clamp capacitor voltage V CL-MAX  and the maximum power rating P MAX  of the floating supply. In that way, if the voltage across the clamp capacitor increases, a higher amount of current will be drawn and vice versa. This provides a balancing effect as the clamp capacitor with the higher voltage will be made to provide a higher amount of current than that with a comparatively lower voltage level. The value of the emulated resistor may be such that the resistive load characteristic overcomes that of the constant power load, so that the total net power has a resistive characteristic. The extra power that is drawn can be dumped into a long term storage (LTS) stage such as  211  in  FIG. 2 . 
     Thus a current I DEMAND  could be drawn according to 
     
       
      
       I 
       DEMAND 
       =V 
       CL 
       /R 
       VIRTUAL  
      
     
     where 
         R   VIRTUAL =( V   CL-MAX ) 2   /P   MAX    
       FIG. 7  illustrates this control strategy being applied for a simulated director switch as discussed above in relation to  FIG. 2 , i.e. a director switch with first and second director switching units, with the first switching unit having a higher starting instantaneous voltage than the second switching unit. Again the simulation covered several power cycles of the converter arm with the same starting conditions. 
     Again the top plot of  FIG. 7  shows the total voltage  701  across the director switch and the voltages across the first  702  and second  703  director switch units respectively. The middle plot shows the voltages of the first  704  and second  705  clamp capacitors and the lower plot shows the current draw of the floating power supplies of the first  706  and second  707  director switch units. It can be seen that the average current demand of the floating power supply of the first director switching unit is greater over the first power cycle than that off the second director switch unit. This reduces the voltage imbalance of the clamp capacitors and over the course of several power cycles the voltage imbalance is effectively removed. 
     In some embodiments the current drawn may additionally be based on the rate of change of the clamp capacitor voltage. For instance the current drawn could be reduced when the clamp capacitor is discharging and reduced by a greater amount if the rate of decay of the clamp capacitor voltage is high. Thus, in one embodiment, the current drawn from the clamp capacitor may be 
         I   DEMAND   =K   1. ( V   CL   /R   VIRTUAL )+ K   2. ( dV   CL   /d   t ) 
     where K 1  and K 2  are weighting factors for weighting the contribution of the resistive and dV/dt characteristics. 
     In some embodiments a resistive load profile may be achieved by controlling the crowbar circuit so as to provide a desired average resistance value. 
     As will be understood by one skilled in the art the crowbar circuit is provided to allow for rapid-discharge of the clamp capacitor after an event that produces an overvoltage across the IGBT, typically a hard-switching event. The crowbar circuit has the crowbar resistor  205  in series with a semiconductor switching element, e.g. transistor  207  for activation of the crowbar circuit. The resistance of the crowbar resistor  205  is chosen so as to provide the desired discharge characteristic. 
     Embodiments of the present invention may operate the crowbar circuit so as to draw a current in a similar fashion as described above. The transistor  207  of the crowbar circuit is controlled so as to enable and disable the current path via the crowbar resistor  205  in a sequence that draws, on average, the required current. In effect the duty cycle of the transistor  207 , i.e. the proportion of time that the transistor is conducting compared to the proportion of time that the transistor is non-conducting, is controlled so that the current path via the crowbar resistor has a desired average resistance. 
     In one embodiment the transistor  207  may be controlled in a pulse-width-modulated (PWM) fashion in a switching cycle with a desired duty cycle. The duty cycle may be controlled to provide a desired average resistance R AV  where 
     
       
      
       R 
       AV 
       =R 
       CB 
       /D  
      
     
     where R CB  is the value of the resistance of the crowbar resistor  205  and D is the duty cycle. The value of the equivalent resistor may be chosen, in a similar fashion as discussed above, so that the equivalent resistor provided by the switched crowbar path draws a maximum rated power P MAX  at a maximum expected voltage of the clamp capacitor V CL-MAX , i.e. such that 
         R   AV =( V   CL-MAX ) 2   /P   MAX    
       Thus  D= ( P   MAX   .R   CB ) V   CL   _   MAX   2    
       FIG. 8  thus illustrates that a controller  801  may be used to control the crowbar transistor  207 . The value of the crowbar resistor R CB  and the desired average resistance value R AV  may be provided to a divider  802  which derives the required duty cycle D, this may be provided to a PWM generator  803  to control switching of the transistor. The PWM generator may for instance comprise a triangle wave generator that generated a repeating ramp waveform based on a switching frequency F SW  and a comparator that compared the ramp waveform to a threshold set based on the duty cycle, although many types of PWM generator are known and may be used. The switching frequency F SW  should be high enough that the action of the equivalent crowbar resistance is sufficiently smooth. A switching frequency of the order of a few hundred Hz may be sufficient. 
       FIG. 9  illustrates this control strategy being applied for a simulated director switch as discussed above in relation to  FIG. 2 , i.e. a director switch with first and second director switching units, with the first switching unit having a higher starting instantaneous voltage than the second switching unit. Again the simulation covered several power cycles of the converter arm with the same starting conditions. In this case the strategy was employed to provide an emulated resistive load of 22 W at 1250V, using PWM control with a switching frequency of 500 Hz and a crowbar resistor of 2.35 kOhm. 
     Again the top plot of  FIG. 9  shows the total voltage  901  across the director switch and the voltages across the first  902  and second  903  director switch units respectively. The middle plot shows the voltages of the first  904  and second  905  clamp capacitors and the lower plot shows the current draw of the floating power supplies of the first  906  and second  907  director switch units. The simulated results show similar performance the results of  FIG. 7 , with just some increase in switching ripple in the voltages and currents. 
     It will of course be appreciated that the voltage balancing strategies discussed above, e.g. operation in a voltage balancing mode, may result in additional power being drawn which is in excess of the rated power demand of the gate driver and auxiliary electronic circuitry. In some embodiments any additional power drawn over and above what is needed for the gate driver and auxiliary electronic circuitry  204  may be used to charge the long term storage  211  of the floating power supply. In some embodiments in the event that any charging of the long term storage  211  is required a control of the director switch unit may be configured to adopt one of the strategies described above to charge the long term storage. In this way any voltage imbalance that is present between the director switching units will be reduced or eliminated during the charge up period of the long term storage. 
     In some embodiments there may further be an additional energy reservoir, e.g. an additional capacitive reservoir. If the long term storage  211  of a power supply of a director switch unit is already fully charged, any additional power drawn by the use of the balancing strategy may be stored in the additional capacitive reservoir. Such an additional capacitive reservoir should have enough capacitance to allow the operation of the balancing strategies for a number of supply cycles. In the event that the additional capacitive balancing reservoir becomes fully charge, it may remain charged, to support any possible loss of supply power. However, if balancing action is required again, it may be rapidly discharged, by dissipating its power through, for instance, a resistive crowbar, before it is used again. Alternatively, the DC-DC converter or the energy storage circuits could be configured so that they have regenerative features. In that case, the stored energy could be returned back into the power system at an appropriate point.  FIG. 10  illustrates a director switch unit having a floating power supply with long term storage  211  for ensuring a source of power for the gate electronics  204  and also an additional balancing reservoir  1001  with a crowbar circuit so it can be discharged when required. The long term storage and balancing reservoir  1001  may be controlled by a director switch unit controller  1002 , which may monitor the voltage of the clamp capacitor  202 . 
     The director switch unit controller of each director switch unit may provide periodic measurements of the voltage of its associated clamp capacitor to a higher level director switch controller.  FIG. 11  illustrates at least part of a converter arm  1100  having a director switch  104  comprising a plurality of director switch units  200 . Each director switch unit may provide periodic measurements of the voltage of the associated clamp capacitor V CL1 -V CLn  to the director switch controller  1101 . Such a director switch controller may determine if there is any voltage imbalance between the clamp capacitors and determine appropriate control signals CNT 1  to CNT n  to control the director switch units to operate in a voltage balancing mode. The voltage balancing mode of operation could be applied to all director switch units, or in at least some instance, only to those that are severely unbalanced. Additionally or alternatively, the balancing mode of operation may be activated on a local basis by a switch unit controller  1002 , for example if the voltage across the clamp capacitor exceeds a high or low threshold window. 
     It will be appreciated that in the embodiments described above the input DC/DC converter of the floating power supply is controlled to behave, over the short term, as a constant power load but the operation may be varied over the longer term, e.g. by varying when the DC/DC converter is active or changing the power level, so as to behave as a resistive load over a longer time scale. In some embodiments it would be possible to instead control the DC/DC converter of the floating power supply to behave over the short term as a resistor, that is to draw a current that is proportional to the voltage across its input terminals. 
       FIG. 12  illustrates such an embodiment where the DC/DC converter  209  of the floating power supply is controlled by a controller  1201 . The voltage V CL  of the clamp capacitor may be supplied to a first control block  1202  that emulates a resistor and determines a suitable current IR for the DC/DC converter to behave as a restive load. The current may be supplied to a current controller  1203  as an input current I IN  for controlling the DC/DC converter  209 . 
     This approach has the drawback that control over the output voltage is lost. However, to overcome that problem, a bulk storage system  1205 , for example a comparatively large capacitor, may be connected to the output of the DC-DC converter  209  to keep the output voltage within upper and lower threshold boundaries. To provide the desired control a voltage monitor block  1204  may monitor the voltage of the storage system  1205 , and hence the output voltage, against lower and upper thresholds (with hysteresis applied). A further voltage monitor  1208  may be arranged to monitor the voltage of the clamp capacitor  202  with respect to a further threshold level. If the output voltage drops lower than the lower threshold level, voltage monitor  1204  will generate a control signal for charge of the bulk storage capacitor  1205  until the output voltage has reached the upper threshold level. However, that charge will only be enabled if the input clamp capacitor voltage is higher than the further threshold as monitored by block  1208 . When both conditions defined by monitoring blocks  1204  and  1208  are simultaneously satisfied, then an extra load current defined by current supply block  1207  will be added to the resistive current demand I R  to be drawn by the DC-DC converter as an input current I IN . If the output voltage gets too high, a crowbar circuit  1206  could be activated to discharge it. 
     In addition to or instead of controlling the floating power supply to behave as a resistive load in some embodiments the resistive loading of the clamp capacitor could be provided by arranging one or more loading resistors to provide the resistive loading. Such loading resistor(s) could be arranged in one or more different locations, for instance across the clamp diode  203 , across the clamp capacitor  202  and/or across the IGBT  110  as illustrated in  FIG. 13A .  FIG. 13A  illustrates a switch unit having a director switch  110  and clamp capacitor  202  and snubber diode  203  as described previously with a floating power supply  208  which, in this example is operated as a constant power load. There is also at least one loading resistor  1301 .  FIG. 1301  shows a loading resistor coupled across, i.e. in parallel with, the snubber diode  203 . 
     In use the clamp capacitor of an individual switch unit will be at a certain voltage level V C  and a certain current is will be drawn from clamp capacitor  202  according to the power requirements. As shown in  FIG. 13B  below a certain voltage level, V 1 , of the clamp capacitor the constant power characteristic of the floating power supply will dominate and the current drawn will vary according to the power demand P and the voltage V C  according to the relationship P/V C . However above the voltage level V 1  there effect of the loading resistor will start to take over and the current drawn from the clamp capacitor will follow the profile resistive loading, i.e. according to V C /R L  where R L  represents the effective resistance of the loading resistors. 
     Embodiments may therefore include one or more loading resistors with appropriate resistance values such that the voltage level V 1  at which the resistive characteristics start to dominate is within the normal expected range of voltages of the clamp capacitors  202  of the switching units. In other words the voltage V 1  at which the resistive effects start to dominate is lower than a nominal operating voltage for the clamp capacitors  202  of the switch unit. 
     As mentioned there may be one or more such loading resistors, for instance a loading resistor may additionally or alternatively be arranged in parallel with the clamp capacitor  202  as illustrated in  FIG. 13A . 
     One skilled in the art will appreciate that the loading resistors would be in addition to or instead of, and will perform a different function to, any resistors typically provided, e.g. resetting the voltage across the clamp capacitor, allowing slow discharge of the clamp capacitor for safety or providing static balancing across the IGBTs. For example as mentioned above some designs of VSC may already include a discharge resistor connected in parallel across the snubber diode  203 . This discharge resistor is intended to allow slow discharge of the clamp capacitor voltage after power down of the system and thus has a high resistance value. Likewise a discharge resistor may be connected in parallel with the clamp capacitor to allow for slow discharge following power off of the system. Again this requires a high value of resistance. For such high value resistances the voltage value at which resistive effects would dominate the constant power load characteristics would be well outside the normal operating voltage of the clamp capacitors. Thus during normal, i.e. non-faulted operation within nominal parameters a conventional switching unit would behave according to the constant power loading only and the value of the discharge resistors would not provide operation in a voltage balancing mode. Likewise a balancing resistor may typically be provided across the switching element  110  to ensure correct voltage balancing across the series connected switching elements in the off state, such a resistor may be chosen to have a relatively high value to provide a current which is only slightly higher than the off-state leakage current of the switching elements  110 . Again the resistance values of the loading resistors will be significantly different to those of the conventional balancing resistors and thus will provide a different loading response. 
     The methods and apparatus described above thus achieves the balancing of the clamp capacitors, even under extreme operating conditions, such as a high constant power load being drawn in the presence of significant circuit mismatch. The techniques described herein also allow recovery of the balancing of the clamp capacitors after they have been upset by an external disturbance. Any corrective action is only taken however when the clamp capacitors are imbalanced, returning to an ideal switching pattern afterwards. This provides a very stable operation in normal conditions. The voltage across the various semiconductor switching elements is thus balanced as a consequence of balancing the clamp capacitors. 
     The various embodiments have been described in respect of an AAC type converter but it will be appreciated that the techniques are applicable to any type of VSC comprising a director switch formed from director switch units having a switching element and a clamp capacitor connected across the switching element where such director switch units also have a floating power supply that draws power from the clamp capacitor. 
     It should be noted that the above-mentioned embodiments illustrate rather than limit the invention, and that those skilled in the art will be able to design many alternative embodiments without departing from the scope of the appended claims. The word “comprising” does not exclude the presence of elements or steps other than those listed in a claim, “a” or “an” does not exclude a plurality, and a single feature or other unit may fulfil the functions of several units recited in the claims. Any reference signs in the claims shall not be construed so as to limit their scope.