Patent Publication Number: US-8527003-B2

Title: System and apparatus for high data rate wireless communications

Description:
RELATED APPLICATIONS 
     This application is a continuation of International Application No. PCT/US2005/040924, which designated the United States and was filed on Nov. 10, 2005, published in English, which claims the benefit of U.S. Provisional Application No. 60/714,393, filed on Aug. 31, 2005, U.S. Provisional Application No. 60/658,018, filed on Mar. 2, 2005, U.S. Provisional Application No. 60/637,076, filed on Dec. 17, 2004 and U.S. Provisional Application No. 60/627,045, filed on Nov. 10, 2004. The entire teachings of the above applications are incorporated herein by reference. 
    
    
     BACKGROUND 
     Wireless communications have become a popular and essential communications medium both nationally and globally. Over the past twenty years, the number of users of Public Land Mobile Networks (PLMN) or cellular telephone networks has grown to over a billion subscribers using networks that provide geographic coverage throughout the world. While these networks primarily provide voice communications, they also enable relatively low rate data communications (e.g., 9.6-140 kbps). 
     As these wireless networks have become more integrated with land-line data networks such as the Internet, the desire and demand to extend the features available within the land-line Internet to wireless devices has increased dramatically. World Wide Web (WWW) access, games, multimedia messaging including pictures and sound, music, enterprise applications, and streaming video are among the service capabilities desired using wireless data devices. Unfortunately, the limited data rates of the supporting wireless networks have resulted in unacceptably slow performance when using these services or other data applications. 
     Recently, PLMN providers have launched CDMA2000 Evolution Data Only (EVDO) networks that provide up to 2 Mbps and Third Generation GSM (3GSM) that provide approximately 300 kbps data rates. Also, the IEEE has standardized Wireless Local Area Network (WLAN) technology including the 802.11 b and 802.11 g standards that provide 11 Mbps or 54 Mbps data rates respectively. The IEEE is working on a new standard referred to as the 802.11 n that will provide data rate greater than 100 Mbps. These newer data standards and networks provide significantly higher data throughput in order to meet the increased demand for wireless data that is needed to enable certain data applications and Internet services within wireless devices. Other wireless data networks also exist including satellite, Specialized Mobile Radio (SMR), private/trunked, Cellular Digital Packet Data (CDPD), fixed wide area networks (WAN), metropolitan area networks (MAN), and personal area networks (PAN) using the Bluetooth protocol or Ultra Wideband (UWB) technology. 
     PLMN networks are generally referred to as cellular networks because they employ a frequency re-use architecture in which wireless access channels are grouped into geographically-located cells and sectors. The size of each cell depends on the output power of the network base station transceiver associated with each cell. Each access channel uses a certain frequency band in one geographic cell that is re-used in another cell, geographically separated from the first cell, by another access channel where the likelihood of interference is minimized. These networks also use a centralized switch or server to enable a wireless device to move from cell to cell while maintaining a persistent data connection. In the United States, cellular and Personal Communications Service (PCS) networks operate in the licensed commercial 824-849 MHz, 869-894 MHz, 901-941 MHz and 1850-1990 MHz. Access data channels, however, are bandwidth limited to 12.5-150 KHz and 25 MHz, depending on the service offered. 
     WLAN networks employ wireless access points that communicate with multiple wireless devices simultaneously via a set of fixed access channels. Typically, these networks use contention protocols such as Carrier Sense Multiple Access Collision Avoidance (CSMA-CA) to enable multiple users to share the same wireless access channels emanating from a transceiver access point. These WLAN networks are generally referred to as wireless Ethernet networks because the access mechanism is similar to conventional Ethernet networks. While WLAN networks may be centrally controlled, they are often used by individuals as wireless bridge or router connections to a local area network (LAN). WLAN networks may operate in the 900 MHz, 2.4 GHz and 5.5 GHz unlicensed bands. The 802.11b is limited to a data rate of 11 Mbps, while 802.11b and 802.11g are limited to 54 Mbps. The 802.11n, which is under development, is expected to have a data rate of greater than 100 Mbps. 
     Two high data rate PAN standards under development are 802.15.3a and 802.15.3c. The former has two competing proposals, namely MB-OFDM and DS-UWB, while the latter is based on the 60 GHz technology. MB-OFDM and DS-UWB have maximum data rate of about 480 Mbps and 1 Gbps respectively at a distance of about 3 meters. The 60 GHz technology is expected to have greater data rate and distance. 
     While these newer wireless data networks provide improved data throughput for users, the desire for even greater throughput is already creating the demand for higher data rate wireless networks that support advanced data application such as high-resolution video conferencing, HDTV connectivity, broadcast video, video-on-demand, online training, distance learning, peer-to-peer collaboration, file transfers, data mining, database applications (e.g., CRM, ERP), and e-mail with attachments. Furthermore, high data rate wireless networks are arguably less costly than cable or copper to install and maintain by an enterprise or user. Concerns regarding reliability, security, quality of service, and coverage of high data rate wireless networks, however, must be addressed. 
     SUMMARY 
     The embodiments of this invention include a wireless network architecture that provides broadband data network coverage over an expandable geographic area, a media access control (MAC) layer to facilitate secure access to the broadband wireless network, a high frequency wireless modem that enables high data rate access to the wireless network, and antenna configurations that enable seamless communications within the wireless network architecture. 
     The invention relates to digital communications modem including a pulse-shaping filter capable of receiving an information signal having a baud rate of at least about 1 gigabit per second. The modem also generates a filtered pulse train that has a reduced bandwidth being substantially less than the baud rate. An encoder receives the filtered pulse train and generating an encoded signal having one of two orthogonal relationships. The modem also includes a combiner receiving two encoded signals having different orthogonal relationships and combining them, with the resulting combined encoded signal having a signal bandwidth substantially within the reduced bandwidth. The modem includes a modulator modulating a carrier signal with the combined encoded signal, such that the spectrum of the modulated signal substantially less than half of the baud rate of the received information signal. 
     Preferably, the modem is combined with a frequency translator that translates the encoded information signal to a millimeter-wave frequency band between about 30 GHz and about 300 GHz. In some embodiments, the frequency of operation is between about 50 GHz and about 70 GHz. 
     The modem can include a pulse-shaping filter, such as a raised cosine filter. The pulse-shaping filter can be implemented as part of a matched filter, the other part of the matched filter provided within a remote receiver. 
     In some embodiments, the encoder includes a dual-rail binary (DRB) encoder. For example, the DRB encoder includes a Hilbert Transformer. The modem can also include a sideband suppressor, such as a filter, that substantially suppresses one of two sidebands of the modulated signal. This can be used with single side band (SSB) or vestigial side band (VSB) modulator. 
     Preferably, the modem uses a first pilot signal source having a center frequency of about half the baud rate. In some embodiments, the modem also uses a second pilot signal source having a center frequency corresponding to the inverse of twice the baud period (i.e., ½ T) at baseband. Preferably, the second pilot signal is provided with a well-defined power level that cam be used for AGC at a receiver. 
     An antenna includes an antenna housing defining a flared aperture adapted to provide an azimuthal beamwidth of at least about 45 degrees. In some embodiments the azimuthal beamwidth is at least 80 degrees, and even 90 degrees. An offset feed port offset with respect to the flared aperture to avoid blockage thereof. The antenna also includes a reflective surface disposed above the offset feed port and behind the flared aperture providing a line-of-sight reflection between the offset feed port and the flared aperture. The antenna provides a far-field gain of at least 21 dBi over a frequency bandwidth of at least about 10% when operating at millimeter-wave frequencies. For example, the reflective surface can be provided with a cosecant-squared shape. In some embodiments, the far-field gain is circularly-polarized having at least about 15 dB of cross-polarization separation. Also, the far-field gain is substantially ripple-free across the azimuthal beamwidth. 
     A linear-to-circular polarization filter includes a dielectric sheet having a thickness of about 3λr/2, λr being the wavelength of an electromagnetic wave propagating within the dielectric. The sheet defines on one side a first series of elongated parallel slots. Similarly, the sheet defines a second series of elongated slots on the opposite side of the sheet, such that the first and second series of slots are substantially aligned with respect to each other. The polarization filter transforms a linearly-polarized electromagnetic wave at one side of the dielectric sheet to a substantially circularly-polarized wave at the other side. Preferably, the linear-to-circular polarization filter is adapted to operate within the millimeter-wave frequency band, when the dielectric material has a relative dielectric constant of less than about 3. 
     A circularly-polarized antenna includes a square-wave input port and a housing coupled at a proximal end to the input port. The housing defines a horn antenna flared along a longitudinal axis, the horn antenna having a cross-shaped aperture. The antenna produces a far-field pattern having an aspect ratio of less than about 3 over a 10% operational frequency bandwidth. Preferably, the antenna is adapted to within the millimeter-wave band, providing a far-field gain greater than about 15 dBi. 
     In some embodiments, a linear-to-circular polarizer is coupled between the input port and a rectangular waveguide, the polarizer converting between circularly-polarized fields at the input port and linearly-polarized fields within the waveguide. The linear-to-circular polarizer can include a rectangular waveguide housing having a first end and a second end, the second end being coupled to the square-wave input port; and a septum disposed within the housing and terminating at the first end only, the septum producing a circularly-polarized field at the square-wave input port responsive to linearly-polarized fields at the first end. 
     An antenna system includes several directional antennas, each antenna providing coverage to a respective portion of the antenna system beamwidth and a switch coupled to each of the several directional antennas, switching on as at least one of the directional antennas to communicate within the response portion of the antenna system beamwidth. 
     A method of wireless networking includes wirelessly communicating network traffic within the millimeter-wave band and applying circular polarization to wireless network transmissions, the circular polarization improving the wireless communications in the presence of multipath. 
     A linear-to-circular polarizer includes a rectangular waveguide housing and a septum disposed within the housing, the septum terminating at a first end of the housing only. The septum partitions a first aperture at the first end of the housing into a first port and a second port. A second end of the housing defines a second aperture providing a third port and a fourth port, the third and fourth ports being distinguished according to the respective linear polarizations of the field at the second aperture. A circularly-polarized field excited at the second aperture produces linearly-polarized fields at the first aperture. 
     The linear-to-circular polarizer device can be operated as a polarization diplexer simultaneously producing linearly-polarized fields at the first aperture responsive to a right-hand circularly-polarized field excited at the second aperture and orthogonal linearly-polarized fields at the first aperture responsive to a left-hand circularly-polarized field excited at the second aperture. Preferably, the polarizer is dimensioned for operation within the millimeter-wave frequency band. 
     A hybrid antenna includes a tapered crossed-waveguide horn having an input port at the narrow end of the taper and a patch antenna coupled to the input port. Electromagnetic energy is allowed to couple between the printed patch antenna and the crossed-waveguide horn. The patch antenna can be circularly polarized, resulting in circularly-polarized radiation. 
     In some embodiments, the crossed-waveguide horn is formed from a metallized injection molded part. The patch antenna can include a dielectric substrate having a conducting patch on a side of the substrate facing the horn. A microstrip feed is disposed on an opposite side of the substrate with respect to the conducting patch, electromagnetic energy coupling efficiently between the microstrip feed and the conducting patch. The hybrid antenna is capable of providing an operational bandwidth of at least about 10% at frequencies within the millimeter-wave band. 
     A wireless networking system provides a communication capability for remote users operating wirelessly within the extremely-high-frequency (EHF) band. The networking system includes a wireless cell defining a geometric shape and a number of wireless access points, each adapted to communicate wirelessly with a remote user within the EHF band using a wireless networking protocol. The plurality of wireless access points can be distributed about the area of the wireless cell and or along its perimeter. In some embodiments, the wireless communications provide a bit error rate of less than 10 −6 . 
     In some embodiments, the geometric shape of a cell is a rectangle, with a respective wireless access points placed at each-of its corners and capable of covering a respective 90-degree sector of the cell. Network extenders can be deployed within the cell between opposing pairs of the plurality of wireless access points to extend the range of wireless access therebetween. Preferably, each of the number of wireless access points is adapted to communicates with the remote user in a channel having a bandwidth of at least 1 GHz. The network can employ a wireless protocol, such as a wireless Ethernet protocol (e.g., IEEE 802.11; IEEE 802.15; and IEEE 802.16). 
     A method for wirelessly networking several remote users using a networking protocol operating within the extremely-high-frequency (EHF) band includes providing a communication channel within the EHF band between at least one of the plurality of remote users and the wireless access point. The communication channel is segmented into a number of frames, each defining a frame marker identifying a respective one of the frames. The remote user requests a channel bandwidth during a bandwidth request phase. The requested bandwidth is allocated during a bandwidth allocation phase, if available. Data is communicated downstream a wireless access point to a remote user using the allocated channel bandwidth. Data is also communicated upstream data from the remote user to the wireless access point using the allocated channel bandwidth. Allocation of the requested bandwidth is accomplished using a network element, such as a management server. 
     The requesting step further includes determining whether the bandwidth request phase is being used by another remote user. If not, the request is made. However, if in use, the requester waits for a period of time before a reattempt is made. The method of claim  60 , wherein the downstream transmissions are variable, depending on traffic. Packet numbers can be assigned to different remote users and the order of packet transmissions can also be altered based on a link parameter. 
     A method of providing a digital communications capability with a remote network user wirelessly within the extremely-high-frequency (EHF) band includes receiving from a remote wireless source a signal. The received signal includes a first pilot signal at a first pilot frequency, a second pilot signal at a second pilot frequency, and modulated data. The frequency of the first pilot signal is determined and used to determining the frequency of the second pilot to a frequency accuracy substantially greater than the first. The second pilot frequency can then be used to detect the modulated data. A signal amplitude of one of the first and second pilot signals can be determined at the receiver and used to adjust gain within the receiver using automatic gain control. Preferably, one of the first and second pilot signals is transmitted at an amplitude known to the receiver. 
     A communication system includes a wireless modem for communicating between a remote station and at least one of a group of wireless access points. A circularly-polarized antenna is coupled to the modem, and a sectorized antenna is provided to limit wireless communications between the remote station and a subset of the plurality of wireless access points. Preferably, the wireless communications operates within the millimeter-wave frequency band. For example, the wireless communications operate between about 50 and about 80 GHz. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The foregoing will be apparent from the following more particular description of example embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating embodiments of the present invention. 
         FIG. 1  shows the network components of an embodiment of the high data rate wireless network; 
         FIGS. 2A-2B  show embodiments of the micro-cell architecture wherein the STA connects with the two strongest WAPs in its vicinity; 
         FIG. 3  shows exemplary frequency assignments of an embodiment in the 60 GHz range; 
         FIG. 4  illustrates the attenuation of a signal of an exemplary mmwLAN embodiment over distance; 
         FIG. 5  shows the multipath effects of wireless transmissions; 
         FIG. 6  shows how mobility is supported in a particular embodiment as an STA moved within a micro-cell; 
         FIG. 7  shows an embodiment of an high capacity cell configuration in which two aggregators support two micro-cells; 
         FIG. 8A  shows an embodiment wherein a single aggregator supports multiple cells simultaneously; 
         FIG. 8B  shows an embodiment wherein multiple cells provide coverage over an extended area; 
         FIG. 9  is a functional block diagram of an embodiment of the wireless data system; 
         FIG. 10  illustrates an embodiment of the Media Access Control (MAC) layer; 
         FIG. 11  illustrates a persistent trace of an oscilloscope configured to display an exemplary “eye” diagram; 
         FIG. 12  shows a Raised Cosine Waveform; 
         FIG. 13  is a block diagram of an exemplary quadrature phase shifter circuit; 
         FIG. 14  illustrates an exemplary Dual-Rail Bipolar Coding waveform; 
         FIG. 15  is a spectral diagram showing the spectra of a Nyquist pulse and a Dual-Rail Bipolar coded signal; 
         FIG. 16  is a spectral diagram showing the resulting spectrum of the coding of  FIG. 15  applied to the Nyquist pulse of  FIG. 15 ; 
         FIG. 17  is a block diagram of SSB demodulator with pilot signal recovery; 
         FIG. 18  shows the location of Pilot  1  and Pilot  2  in the frequency domain to enable VSB demodulation; 
         FIG. 19  is a block diagram of SSB or VSB demodulator using two pilots; 
         FIG. 20  shows the location of Pilot  1  and Pilot  2  in the frequency domain to enable SSB demodulation; 
         FIG. 21  illustrates the Q characteristics of an exemplary notch filter; 
         FIG. 22  shows the output magnitude when a notch filter is place in parallel with a circuit load; 
         FIG. 23  illustrates magnitude and phase response of two notch filters combined in parallel with each tuned adjacent to a respective side of a center frequency to reduce the sensitivity of the notch filter; 
         FIG. 24  is a block diagram of an embodiment of a high data rate transmitter using Dual-Rail Bipolar, Single Side Band and dual pilots; 
         FIG. 25  is a block diagram of an embodiment of a high data rate receiver configured to receive a Dual-Rail Bipolar, Single Side Band having dual pilots; 
         FIG. 26A  is an exemplary circuit implementing the SRN filter of the transmitter of  FIG. 24 ; 
         FIG. 26B  is an exemplary additional circuit that couples to the circuit of  FIG. 26A ; 
         FIG. 26C  is an exemplary group delay equalizer that can be used in the receiver of  FIG. 25 ; 
         FIG. 26D  is an exemplary Hilbert Transformer of the transmitter of  FIG. 24 ; 
         FIG. 27  is an exemplary response of the receiver of  FIG. 25  to a logical 1 transmitter from the transmitter of  FIG. 24 ; 
         FIGS. 28A  and B illustrate an exemplary WAP antenna configuration and elevation patter gain, respectively; 
         FIG. 29A  illustrates an exemplary linear-to-circular polarizer; 
         FIGS. 29B  and C illustrate response of the polarizer of  FIG. 29A  to an incident E-field; 
         FIGS. 30A  and B illustrate an exemplary high-frequency linear-to-circular polarizer; 
         FIGS. 31A-31D  illustrate simulated far-field gain patterns for an exemplary WAP antenna at three different frequencies, respectively; 
         FIGS. 32A-32C  illustrate simulated far-field gain patterns for an exemplary WAP antenna combined with the linear-to-circular polarizer of  FIG. 30B ; 
         FIGS. 33A  and B illustrate aspect ratios at different frequencies for the exemplary WAP antenna-polarizer combination of  FIGS. 32A-32C ; 
         FIG. 34  is a perspective illustration of an embodiment of an exemplary WAP antenna-polarizer combination; 
         FIGS. 35A and 35B  are elevation patterns for different embodiments of a WAP antenna-polarizer combination; 
         FIG. 36A  is an end view of a cross-shaped, square waveguide horn antenna; 
         FIG. 36B  is a cross section of the waveguide horn antenna of  FIG. 36A ; 
         FIG. 37A  is a perspective diagram of a septum polarizer; 
         FIGS. 37B and 37C  show the polarization performance of the polarizer of  FIG. 37A  responsive to E and H fields; 
         FIG. 38  shows the antenna gain and aspect ratio of a right-hand-circularly polarized cross-shaped, square horn antenna; 
         FIGS. 39A  and B show exemplary far field antenna patterns for the antenna of  FIG. 36A  for right-hand circular polarization and left-hand circular polarization, respectively; 
         FIG. 40A  shows a septum polarizer combined with a cross-shaped horn antenna; 
         FIG. 40B  shows a cross section of the antenna of  FIG. 40A ; 
         FIGS. 41A-41C  illustrate the gain and aspect ratio for the antenna of  FIG. 40A  at three different frequencies, respectively; 
         FIG. 42  illustrates an end view of a horn antenna fed with a microstrip patch antenna feed; 
         FIG. 43A-43C  illustrate circularly polarized patch antennas having an inverted microstrip feed; 
         FIGS. 44A-44B  illustrate a suspended circularly-polarized patch antenna with an inverted microstrip feed; 
         FIG. 44C  illustrates the return loss performance of the device of  FIGS. 44A and 44B ; 
         FIGS. 45A-45H  illustrate exemplary performance parameters of the device of  FIGS. 44A and 44B ; 
         FIG. 46B  shows various views of a standard microstrip to inverted microstrip transition; 
         FIG. 46A  shows the return loss of the device of  FIG. 46B ; 
         FIGS. 47A and 47B  show an exemplary suspended patch antenna with inverted microstrip feed combined with a cross-shaped, square horn antenna; 
         FIGS. 48A through 48F  illustrate exemplary performance parameters of the device of  FIGS. 47A and 47B ; 
         FIGS. 49A and 49B  are antenna patterns for an exemplary WAP antenna; 
         FIG. 50  is an exemplary STA antenna and its corresponding elevation pattern; 
         FIG. 51  is cross section of an exemplary WAP extender used through a wall; and 
         FIG. 52  is an exemplary link configuration comparing performance link performance with and without a WAP extender. 
     
    
    
     DETAILED DESCRIPTION 
     Certain embodiments of this invention include a wireless network architecture that provides broadband data network coverage over an expandable geographic area, a media access control (MAC) layer to facilitate access to the broadband wireless network, a high frequency wireless modem that enables high data rate access to the wireless network, and antenna configurations that enable seamless communications within the wireless network architecture. 
     Wireless Network Architecture 
     One embodiment of this invention features a wireless network  100  that reliably, securely, and efficiently provides high data rate connections for wireless devices. In the embodiment of  FIG. 1 , several Wireless Access Ports (WAP)  110   a ,  110   b ,  110   c ,  110   d  and wireless devices  115   a ,  115   b ,  115   c , e.g., mobile stations (STA) connected to or integrated within a laptop or desktop personal computer (PC)  120 , utilize 60 GHz wireless modems to exchange digital information within a millimeter wireless LAN (mmwLAN)  100 . An aggregator  125  aggregates network traffic to and from each WAP  110 . The aggregator  125  interfaces with a Network Management Server  130  via a router  135  and the Internet or a local Intranet  140 . Because high frequency communications are susceptible to rapid attenuation, the wireless network  100  may utilize Network Extenders  145  to extend the propagating distance of WAP or STA signals through objects such a walls  150 ′,  150 ″. Further details regarding the wireless network embodiment are described as follows. 
     Referring to  FIG. 2A , the square-shaped (or rectangular-shaped) communications cell  200  utilizes four WAPs  210   a ,  210   b ,  210   c ,  210   d  dispersed to each corner of the cell  200  with each assigned a unique frequency band according to  FIG. 3 . Each frequency band occupies a respective  695  MHz channel with each channel producing a data rate of 1.2 Gbps. A unique color code reference can be designated for each WAP  210  to illustrate the frequency diversity within each cell. An exemplary MMW spectrum allocation for Canada and the U.S. is illustrated in  FIG. 3  for the 57-64 GHz band. Thus, the blue, red, yellow, and green color-coded bands occupy the 57.745-58.440, 59.135-59.830, 60.525-61.220, and 61.915-62.610 GHz frequency spectrum respectively. Additionally, in one embodiment, the WAP  210  associated with each red, green, blue, and yellow band is located in the same location within a cell as illustrated in  FIG. 2A . The four-WAP arrangement may be implemented at any frequency subject to the propagation characteristics at that frequency. The size of the coverage region can be increased by using additional WAPS by ensuring that adjoining WAPs operate at a different channel. 
     A STA  215  operator within the cell  250  can receive signals from the four WAPs  210   a ,  210   b ,  210   c ,  210   d . In some embodiments, the STA  215  identifies the best two WAPs  210  and communicates to those two (e.g., at 2 to 4 Gbps). By illuminating a cell  200  into multiple frequencies (i.e., colors) provides the resulting network with frequency and space diversity.  FIG. 2B  illustrates a 12-WAP  210  configuration. The WAPs  210  can be distributed about a perimeter of the cell  200 ′. Each of the WAPs  210  transmits on a respective one of a number of channels. For example, four channels (or colors red, green, blue, yellow) are used among the  12  WAPs  210 , some of the WAPs  210  using the same channels. 
     A STA  214  operating within the cell  200 ′ provides sectorized-azimuthal coverage. For example, the STA  215  includes eight sectors  250   a - 250   h  covering 360° in azimuth. 
     Based on the attenuation characteristics illustrated in  FIG. 4  for a system operating at approximately 60 GHz, each WARP  210  can be positioned approximately 50 meters from its adjacent WARP  210  along the perimeter of a square cell  200 ′. This results in at least 20 dB margin given a STA antenna gain of 15 dBi, a WAP antenna gain of 24 dBi and an input power of 16 dBm In other embodiments utilizing different frequency spectrum, the optimal cell size may vary depending on the signal attenuation characteristics for that spectrum. A WARP  210  is a wireless access point functioning as a layer-2 device such as a network bridge to a LAN or the Internet. Each WARP  210  may be connected to an aggregator  225  using CAT-5e or CAT-6 cabling, or fiber  220 . An aggregator  225  is typically a layer-2 device that manages the Stations (STA)  215   a ,  215   b , removes duplicates packets, and load balances the traffic. At the same time, the STA  215  also interfaces with a network router  235 , management server, or gateway. The aggregator  225  may utilize any standard network protocol such as Ethernet and interface with any network including a fiber-based network. 
     Each STA  215  may be a mobile or stationary client device that may be integrated within or interface with another device, such as a laptop, a PC, a television, a set-top box or home entertainment equipment, a printer, a server, a Personal Digital Assistant (PDA), a wireless telephone, or a Voice-over-IP (VoIP) device. Typically, the STA  215  scans its immediate area to identify the two strongest WAPs  210  within its vicinity. In one embodiment, the STA  215  determines which two WAPs  210  within a particular cell that provide the strongest two signals and initiates communications with both WAPs  210 . In another embodiment, the aggregator  225 , a central server, or another network element may designate which WAPs  210  can communicate with a particular STA  215 . 
     Data can be transmitted and received to-and-from one or more WAPs  210  either simultaneously or in different time slots. Consequently, the probability of receiving any error at the STA  215  is significantly reduced. For example, the probability of losing one packet with such a configuration is typically 0.1% while the probability of losing two packets is reduced to approximately 10 −6 . 
     The network architecture requires placement of WAPs along the periphery of a 52.5 m×52.5 m region. The region is populated by stations (STAs) which have 8-sectorized beams. These STN antennas improve the outage probability by more than two orders of magnitude when compared to omni-directional ones. This improvement is attributed to the higher gain of the STA antennas and reduced multipath reflection delay spread y virtue of narrow beam width and circular polarization. A STA can receive multiple signals, each at different frequency, from multiple WAPs. The STN can transmit duplicate packets sequentially to the same or different WAPs in the same or different frequencies. This combination of space, frequency and time diversity mitigates fading and interference and improves reliability and availability. 
     Detailed analysis indicates that the outage probability in a 52.5 m×52.5 m region with four WAPs in each corner is guaranteed to be less than 10 −4  which translates to an availability of 99.99% minimum. The analysis considers  109  randomly placed obstacles, each obstacle having a mean loss of 6 dB and a standard deviation of 3 dB. 
     When twelve WAPs populate the periphery of a 102 m×102 m region, the outage probability drops to less than 10 −5  or 99.999% availability minimum. This scenario  10  considers  436  randomly placed obstacles with 6 dB and 3 dB of mean loss and standard deviation respectively. 
     An office environment with human traffic is demanding. This is because human shadowing loss tends to be around 20 dB. The analysis considered 99 m×99 m region with 192 humans randomly placed and twelve WAPs located along the periphery. The outage probability was less than 10 −2.7  or availability of 99.8% minimum. However, when the STA retransmits the signal, time diversity, the outage probability improves to less than 10 −5  or availability of 99.99% minimum. 
     Because the selected WAPs  210  operate at different frequencies, are positioned in different locations relative to the STA  215 , and are likely different distances from the STA  215 , the STA  215  utilizes the combination of frequency, space, and time diversity to ensure that a wireless connection is not lost or corrupted, resulting in a high degree of network availability. For instance, diversity typically reduces the impact of errors caused by interference due to, among other causes, the multipath effect illustrated in  FIG. 5 . Multipath refers to reflected versions of a transmitted signal arriving at a receiver with different respective delays due to the different path delays. Generally, such multipath is an undesired effect and represents noise at a receiver. 
     At least one way to reduce the effects of multipath is to use a sectorized antenna. A sectorized antenna communicates within a preferred range of directions and ignores directions out side of that range. Such operation will avoids all multipath that does not fall within the preferred range of directions. Another approach to reduce the effects of multipath is to use circularly polarized transmissions. As illustrated, odd numbers of reflections will change the sense of the polarization at a receiver. Thus a receiver configured to receive a right-hand polarized signal will ignore left-hand polarized reflections. Advantageously, sectorization can be combined with circular polarization for even greater performance enhancements. 
     Such a combination of diversity provides greater than 99.999% availability which may exceed commercial grade fiber or CAT 6 availability used as a backhaul for the wireless network. If only one WAP  210  is available, however, an STA  215  may interface with a single WAP  210  even though the probability of losing packets may increase. 
     Depending on modem capabilities and encoding techniques used which are discussed in more detail further herein, STA  215  data rates may reach 3.6 Gbps within a particular cell  200 . Because each WAP  210  typically has a 1.2 Gbps channel, each cell  200  may support between 1.2-12 Gbps data rates depending on the number of WAPs  210  operating with a cell. 
       FIG. 6  shows the WAP  210  handoff process as a mobile STA  215  moves in a clockwise direction within a particular cell  200 . As the STA  215  moves, it constantly monitors the WAPs  210  ( four color-coded WAPs  210  in this case) to determine the signal strength. Based on the signal strength information and based on the instructions from the aggregator  225 , the STA  215  will communicate with one or multiple WAPs  210 . As illustrated in  FIG. 6 , the STA  215  begins in the start position connected to the yellow and green WAPs  210   d ,  210   a , progresses around the cell  200  while connecting to the green and red WAPs  210   a ,  210   b  or the red and blue WAPs  210   b ,  210   c , and eventually reconnects to the yellow and green WAPs  210   d ,  210   a  in the end position. The STA  215  may use a drop and connect technique to switch WAPs  210  or acquire the destination WAP  210  before dropping the carrier of the overpowered WAP  210 . 
     While the coverage area of the exemplary cell  200  of  FIG. 6  is limited to 250 m 2 , the capacity and coverage area may be expanded by simply adding additional cells adjacent to each other. In  FIG. 7  is shown an expanded cell architecture having two cells  200 ′,  200 ″ placed adjacent to each other to double the coverage area of the millimeter wave LAN. Because of the color-coded frequency diversity, the likelihood of interference between first and second cells  200 ′,  200 ″ is minimized. For instance,  FIG. 7  shows that the green and red WAPs  210   b ′,  210   a ′ of the first cell  200 ′ are adjacent to the yellow and blue WAPs  210   c ″,  210   d ″ of the second cell  200 ″. Also, the transmitters and receivers of the green and red WAPs  210   a ′,  210   b ′ of the first cell  200 ′ do not point in the direction of the transmitters and receivers and the green and red WAPs  210   a ″,  210   b ″ of the second cell  200 ″. This configuration allows the same frequencies to be re-used within each cell  200  to greatly increase the overall network capacity while minimizing possible interference caused by using the same frequencies in each cell  200  or cell area.  FIG. 7  also shows that multiple aggregators  225 ′,  225 ″ may coordinate with each other to handle the WAP operations within multiple cells  200  and the increased aggregate data throughput of the system  700 . 
     In  FIG. 8A , the number of cells are expanded to four  800   a ,  800   a ,  800   c ,  800   d  and area of coverage is expanded to 10,000 m 2 . Depending on the throughput capability of the aggregator  825 , the wireless network  805  may maintain a 2-4 Gbps data rate per cell  800  due to frequency re-use among all cells  800  or an overall 2-4 Gbps data rate among the four cells combined. 
       FIG. 8B  provides a further example of a nine cell  800   a - 800   i  configuration that provides coverage over a much wider area to possibly support networking throughout a corporate or college campus. Each of the cells includes four WAPs  810  at its respective corners. Such extensive coverage is advantageous in providing outdoor coverage. Again, the color-coded frequency/WAP configuration within each cell continues to reduce possible interference between cells regardless of the number of cells. It is important to note that the number of cells and possible coverage area may be unlimited due to the frequency re-use and color-coded diversity technique employed within each cell. 
       FIG. 9  is a block diagram that illustrates the components that are common to the WAP  210  and STA  215  elements. In particular, the components include processors  900  providing a Media Access Control (MAC) layer that facilitates access to the broadband wireless network, a high frequency wireless modem  905  that enables high data rate access to the wireless network, a front end  910  that may provide gain and/or filtering and transmit and receive antenna design and configurations  915 ,  920  that enables signal propagation between network elements to form a novel seamless wireless communications system. Each of the components are now described in further detail. 
     Media Access Control 
     In certain embodiments as shown in  FIG. 10 , a Media Access Control (MAC) layer may be employed to coordinated STA access to the WAPs and millimeter wave LAN network. To enable access control, each frequency band or channel, e.g., yellow, blue, red, or green WAP-to-STA 1 GHz channel, may be segmented into frames  1000  with each frame  1000  further including a Frame marker  1010 , Bandwidth Request Phase  1015 , Downstream data  1020 , Bandwidth Allocation Phase  1025 , and Upstream data  1030 . The WAP  210 , aggregator  225 , or another network element typically exercises control as opposed to allowing an STA ad-hoc mode, however such mode may also be allowed. 
     The Frame marker  1010  may be a delimiter field that identifies the beginning of a frame  1000 . The Bandwidth Request Phase segment  1015  may include STA  215  requests for data access. Because multiple STAs  215  may compete for access to the same channel, a contention protocol scheme such as CSMA-CA or the like that uses binary exponential back-off may be employed to ensure that each STA  215  has access to the Bandwidth Request Phase segment  1015 . In other words, STA  215  request bandwidth in contention slots. If the slot is in use, a requesting STA  215  will wait a certain period of time based on an exponential back-off until attempting to transmit to the WAP  210  using that Bandwidth Request Phase segment  1015 . The STA  215  may include Quality-of-Service (QoS) information in the request message. 
     The Bandwidth Allocation Phase segment  1025  may enable a WAP  210  to assign upstream data segments, downstream data segments, and their associated packet numbers to a particular STA  215 . In other words, the WAP  210  may use the Bandwidth Allocation Phase segment  1015  to assign each connected STA  215  a unique slot within the carrier or frequency band data stream. In some embodiments, the WAP  210  assigns packet numbers to different STAs  215  for upstream transmissions. Depending the QoS, type of traffic, or other data criteria, the WAP  210  may use a scheduler to flexibly control the order of data packets sent to each STA  215 . 
     Downstream data, i.e., data sent from a WAP  210  to one or more STAs  215 , is typically delivered in the Downstream data segment  1020 . The WAP  210  delivers data to the STAs  215 . In some embodiments, the WAP  210  delivers data to the STAs  215 . In some embodiments, the WAP provides variable-length downstream transmissions depending on traffic. Depending on the type of data traffic, the size of each downstream data packet may be varied to improve overall throughput efficiency. Downstream data may be transmitted contiguously. Upstream data, i.e., data sent from a STA  215  to the WAP  210 , may be transmitted by a STA  215  according to a unique Upstream data segment slot assigned by the WAP  210 . 
     Logically, the Bandwidth Request Phase segment  1015  is considered an upstream control channel that enables a STA  215  to request access to the network. Also, the Bandwidth Allocation Phase  1015  is considered a downstream control channel that enables a WAP  210  or other network controller to assign upstream and downstream data channels or segments to a particular STA  215  to enable network data exchange. While the upstream and downstream control channels may primarily handle initial STA  215  access to the network, bandwidth requests for existing data sessions may be enabled by allowing the STA  215  to append or “piggy back” request fields onto upstream packets in a previously-assigned upstream data segment or slot. The logical control channels may also support other essential network features such as STA  215  and/or network authentication or key distribution for subsequent authentication or data encryption. 
     For any wireless network where information is inherently vulnerable to interception, security is considered an important feature to prevent unauthorized access to the network or disclosure of information. Certain embodiments provide enhanced security features such as mutual authentication to prevent unauthorized STA access or to prevent a false WAP from attempting to access an STA. Private key, password, Public Key, Public Key Certificates, or Biometrics are among the mechanisms that may be used to enable STA authentication or mutual authentication between the STA and WAP or another network authentication element such as a RADIUS Authentication, Authorization, and Accounting (AAA) server. 
     Certain embodiments may support Extensible Authentication Protocol (EAP) or, more particularly, EAP over LAN (EAP-OL) or other 802.1x security mechanisms. Furthermore, secure and efficient cryptographic key distribution based on Kerberos, PKI, smart cards, Public keys, secret keys, tokens, or manual distribution, or other protocols such as IPsec, SSL, EAP or 802.1x may be employed to enable both authentication and data encryption. Block ciphers such as the Data Encryption Standard (DES), in any of its modes, may be used to encrypt control channels, data channels, or both channels. Publicly known or proprietary symmetric encryption algorithms including IDEA, Triple DES, RC4, RC5, AES (Rijdael), or the like may provide data privacy. Also, public key algorithms such as RSA may be used for privacy while Diffie-Hellman Key Exchange may be used for key distribution. 
     Encryption, scrambling, and other privacy techniques may be employed at multiple locations and network layers simultaneously to enhance network security. For example, AES encryption may be employed between the STA  215  and WAP  210  to provide data link protection over the air. Also, each WAP  210  may uniquely scramble signals on a per channel basis by assigning a unique seed to a STA  215  via the control channel. This per channel digital channel scrambling may be refreshed as often as desired by the WAP  210  depending on the per-channel policies regarding the security of certain users. 
     For example, the WAP  210 , aggregator  225 , or another network security element that supervises network access control may designate varying degrees of security depending on a user&#39;s privileges. Depending on the allowed privileges or sensitivity of access, the WAP  210  may periodically authenticate access, re-seed the access channel scrambling, and perform a new key exchange for AES encryption. Furthermore, the digital channel scrambling may be customizable to enable a network provider to incorporate a preferred scrambling or encryption technique. 
     In certain embodiments, such as when the frequency bands are in the 60 GHz range, techniques may be employed to limit the propagation of signals or confine the energy of such signal to limited areas, thereby reducing the possibility of interception or jamming. Other coding techniques such a code division multiple access (CDMA) may be employed to further prevent signal interception and jamming. 
     While most network MAC functionality may be employed at each WAP  210 , it may also be employed at the aggregator  225 , network router  235 , gateway, or other network server such as a Remote Authentication Dial In User Service (RADIUS) Authentication, Authorization, and Accounting (AAA) server. 
     Wireless Modem 
     Special Requirements of the Millimeter Wave Band Modem 
     While the wireless network architecture and MAC features may support wireless networking at any range of frequencies, a wireless modem embodiment is described herein that supports wireless digital data communications between the STA and WAP in the mmwLAN frequency range, i.e., 50-90 GHz range. It should be noted, however, that the techniques employed to enable cost-effective modulation and demodulation in the mmwLAN range may also apply to wireless modem embodiments operating at other frequencies. 
     In order to produce data rates aggregating to over 10 Gbps over the millimeter wave bands at 50, 60, 70, 80 and 90 GHz which have bandwidths from 2 to 7 GHz, the modem needs to generate a bit rate density around 2 bps/Hz. To produce individual sub-channels of 1+Gbps, bands of least 500 MHz must be used per channel. The Nyquist sampling rate for such a bandwidth would exceed 1 gigasample-per-second. If converted by an A/D converter to 6-bits per sample, this would result in a bit rate of at least  6  Gbps which must be signal processed by a DSP. Such a signal processing architecture is well beyond the current state of the art. The modem proposed here uses Analog Signal Processing (ASP) techniques to accomplish this requirement. 
     Direct frequency synthesis from a crystal source produces a clean, stable signal. Unfortunately, frequencies for the millimeter wave band cannot be synthesized directly with a crystal. One has to begin with a lower frequency oscillator typically based on field-effect transistor (FET) technology or using a low-frequency crystal and then multiply the frequency up to the millimeter wave band in use. The phase noise performance resulting from such oscillators is not as good as can be achieved by a crystal oscillator alone since the process of multiplication also multiplies the phase noise by the same factor. Hence the currently-attainable millimeter-wave frequencies generated have a significant amount of phase noise. Consequently, design of the modem must use phase information with extreme care or it will only be able to achieve reasonable performance at high-received signal power levels. 
     The amplitude of the millimeter wave oscillator and the Local Oscillator (LO) leakage of the millimeter wave up-converter generally cannot be characterized very well and will drift in time. This makes the amplitude and DC in the IF (or baseband) modem signal unpredictable and susceptible to drift. If the modem signal has a DC component, DC wander would lead to errors. Slow Automatic Gain Control (AGC) and DC wander correction loops can be used to mitigate these effects in point-to-point modems. However, in a point-to-multipoint modem, one has to quickly turn the modem around to listen to different source for the duration of a packet, which may last a few microseconds. Hence slow AGC and DC wander correction circuits cannot be employed. The modem must, by design, deal with these problems. 
     The output power amplifier at the transmitter is one of the most expensive components used in a millimeter wave band modem and its cost grows exponentially with the power it has to deliver. Unfortunately, in order to keep the cost in line one has to drive the amplifier as hard as reasonably possible. This produces distortion in the signal in the form of compression. The modem design has to take such distortion into account. Most traditional modem designs require very good linearity in the power amplifier typically obtained by operating the amplifier at a power level that is reduced by 6-10 dB from the 1 dB compression point to keep the amplifier in the linear region. 
     To achieve the high bit rate density, the modulated IF and RF signals must preserve bandwidth by using a spectrally-efficient modulation, such as Single Side Band (SSB) or Vestigial Side Band (VSB) techniques. Circuits that are used to implement these techniques, however, are increasingly inaccurate at the band edges also as the operational bandwidth is increased. For these reasons the bandwidth is minimized and the energy at the band edges are reduced to as low a value as possible. 
     Lastly, the cost of ASP techniques (unlike DSP techniques) rises exponentially with complexity. For example, building adaptive equalizers using ASP techniques is very difficult, expensive and not reasonable (compared to DSP) beyond relatively simple cases. As another example, achieving clock recovery using an adaptive Costas loop results in fairly hard-to-implement circuits that drive up the cost. So the modem is preferably restricted to using simple methods for all the sub-functions. 
     Design of the Modem 
     The modem described here processes sub-channels of bandwidth exceeding 460 MHz to produce data rates beyond 1 Gbps. Thus the resulting bit rate density is approximately 2.2 bps/Hz. The baseband signal has substantially no energy at DC and again at half the baud rate. This allows the receiver to be AC connected thereby eliminating the need for DC wander correction circuits. It also allows for the injection of two low level pilots into the signal: one at DC; and the other at half the baud rate. The DC pilot maps into the sub-carrier frequency of the millimeter wave sub-band being employed. It undergoes the same phase change (including phase noise) as the modem signal. So it can be used to coherently demodulate the signal at the receiver either directly or more likely through an IF stage. Such demodulation allows one to considerably mitigate the phase noise effects as well. 
     The second pilot allows for using a rapid AGC at the receiver, since the second pilot&#39;s amplitude is well controlled at the transmitter relative to the signal. Also, when the coherent demodulation described above, is finished, this second pilot moves to the half-baud rate frequency as defined at the transmitter end. This allows the receiver to achieve very simple clock recovery. Since the second pilot undergoes the same phase translations as the signal, the rising and falling edges of the second pilot can be used for data recovery as well. In essence, the differential frequency between the two pilots allows us to bring transmitter timing to the receiver in spite of the frequency inaccuracy and the phase noise of the millimeter wave oscillators at the transmitter and the receiver. 
     This modem design essentially is a baseband scheme and does not use the phase information. It is also highly resistant to compression and can easily operate through 1 dB compression with minimal eye closure. While this is not intuitively obvious, it is easily verified by simulations. 
     In conclusion, the modem uses a coding technique described herein and called Dual-Rail Bipolar (DRB) combined with Single Sideband (SSB) transmission to achieve the high bit rate density and the spectral nulls at DC and half the baud rate where the dual pilots are inserted. DRB is a novel variant on the traditional technique called Bipolar Coding (or Alternate Mark Inversion (AMI)) used by T-carrier lines deployed by the phone company. The coding scheme allows for use of the dual pilot scheme described above. In concert, these three techniques (DRB, SSB, Dual-Pilot) can be used to meet all of the requirements outlined in the previous section. 
     Wireless Modem: Details 
     Wireless data communication systems such as IEEE 802.11, CDMA 2000, and 3GSM use digital communications as opposed to earlier AM or FM radios that rely solely on analog communications. Digital communications enable the transfer of information in the form of discrete 1s and 0s which usually correspond to approximately 5 volts and 0 volts respectively in a CMOS electronic circuit, e.g., Return-to-Zero (RZ) signals. The transmission of an information sequence results approximately in square waves with approximately 5-volt peaks and 0-volt valleys. Unfortunately, square waves have infinite bandwidth that results in time domain spreading of each square wave when filtered by the finite bandwidth of most real life channels. 
     While it is acceptable to transfer information as binary 1s and 0s, or square wave pulses, in an electronic circuit with essentially infinite bandwidth, Nyquist showed that, in a bandwidth limited environment (e.g., a wireless access channel), infinite bandwidth square wave pulses, after channel filtering create inter-symbol interference (ISI) whereby, due to time domain spreading, the energy of preceding square waves wash over the energy of subsequent square waves (See “Certain Topics in Telegraph Transmission Theory,” Harry Nyquist, Trans. AIEE 47: pp. 627-644, 1928). 
     Thus, a receiver detects the combination of many symbols simultaneously, preventing retrieval of the original information. In other words, the received signal may be undetectable or appear to be noise. 
     In a wireless digital communications environment, information is propagated using electromagnetic energy in the form of carrier waves of a certain wavelength or frequency that are frequency or amplitude modulated with digital information. Modulation is generally a three-step process including: (i) baseband modulation where the binary information bits are shaped to minimize ISI in the limited spectrum; (ii) passband modulation where the baseband signal is further modulated to an intermediate frequency (IF) carrier; and (iii) The IF signal is modulated to the millimeter wave sub-band the signal is supposed to operate in, the radio frequency (RF) signal. 
     A wireless modem is used to both modulate carrier signals with outgoing baseband digital information and to demodulate received signals to recover incoming baseband digital information. Pulse amplitude modulation (PAM) is the most common baseband modulation technique wherein a sequence of information pulses is amplitude modulated into time-translated pulses. PAM is a form of linear modulation in which baseband data bits modulate an in-phase, i.e., cosine carrier. Other forms of modulated PAM include quadrature amplitude modulation (QAM) where doubling of the transmitted data rate is obtained by modulating the quadrature, i.e., sine carrier with independent data. Yet another example is phase shift keying (PSK) where multiple data bits can be coded into one of multiple phases of a carrier. 
     To eliminate the problem of ISI, Nyquist proposed that digital information signals be transmitted using band limited pulses of a form that create no ISI. Nyquist showed that such pulses have a bandwidth somewhat greater than ½ T where T is the baud period. The excess bandwidth is described by a factor α and the overall bandwidth of the pulses is (1+α)/2 T. He showed that when properly designed, such pulses can be made to have a value of 1 at some time t* and undergo periodic zeros every T seconds before and after t*. It is clear that such “Nyquist” Pulses will produce no ISI at times t*+nT where ‘n’ is the set of integers. The only pulse at that time with a non-zero value will be the pulse carrying the n th  data bit (this pulse will have a one value). 
     By using an oscilloscope in persistent mode, the “eye” diagram according to  FIG. 11  may be used to measure the amount of ISI within a signal. Overlapping traces  1100  of the signal produce an “eye” pattern. The width of the one of the openings corresponds to a timing allowance and the height or openness of one the eyes corresponds to an amplitude allowance. For a Nyquist pulse train, the “eye” diagram will appear well-defined and clean. For non-ideal pulses, the “eye” diagram will be less open with the horizontal (timing) and vertical (amplitude) space will close compared with the ideal pulse. The horizontal width of the “eye” diagram can be controlled using an equalizer to compensate for channel dispersion/distortion. The vertical width or height of the “eye” diagram is generally controlled by a pulse shaping filter and amount of noise in the communications channel. 
     Although it is impractical to achieve the Nyquist pulse in real systems, other types of pulse waveforms may be implemented using a pulse-shaping filter to minimize ISI. The raised cosine (RC) pulse is popular as it represents a Nyquist pulse with the shape of an offset cosine pulse that oscillates and decays rapidly.  FIG. 12  illustrates an RC pulse with 50% excess bandwidth. The RC pulse also has zero crossings at multiples of T that eliminates ISI. The RC pulse&#39;s bandwidth may be varied between the minimum Nyquist bandwidth, R/2, and R where R is the symbol rate. As explained earlier, the amount that the bandwidth exceeds R/2 is known as the Excess Bandwidth a factor. The RC pulse width may be decreased by increasing the excess bandwidth. Typically, an RC pulse with 25 to 50% excess bandwidth is employed. The impulse response of an RC is defined as: 
     
       
         
           
             
               
                 
                   
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     Typically, to implement the RC response, RC filtering is split into two parts, with one part at the transmitter and the other at the receiver, to create a matched set of filters. When the RC filter is split into two parts in this manner, each part is known as the Root Raised Cosine (RRC) because the RRC is the square root of the RC frequency response in the frequency domain. When the two filter parts are combined in series at the transmitter and receiver, the result is the original RC filter. The advantage of using a RRC filter is that received information pulses have a low pass response that allows the data information to pass while attenuating high frequency noise. The matched filter pair also correlates the received signal with transmit pulse shape to improve the signal-to-noise ratio (SNR). Such RC filters are typically realized as Finite Impulse Response (FIR) filters in a DSP based system. 
     PAM also employs line coding such as RZ, Non-Return-to-Zero (NRZ), four level, Bipolar (Alternate Mark Inversion (AMI)), or Manchester coding to produce desired spectral properties in the modem signal such as eliminating a Direct Current (DC) component or to creating spectral nulls at desired frequency points. 
     In a PAM digital communications system, the date rate (DR) is proportional to the symbol rate and number of modulation levels. 
     
       
         
           
             
               
                 
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     Using QAM, the number of bits/symbol may be increased, allowing the data rate to be doubled for a given bandwidth or symbol rate by modulating one DR worth of data on a “cosine” carrier and modulating another DR worth of data on a “sine” carrier in quadrature to the original “cosine” carrier. With QAM, a constellation of data points is created based on phase and amplitude. For example a 4-QAM constellation will have 4 points, each point associated with two data bits, while a 16-QAM constellation will have  16  -points, each point associated with 4 data bits. Because QAM increases the number of data points, the distance between these data points decreases resulting in a reduction of signal-to-noise ratio (SNR) at which satisfactory performance can be obtained. However, compared to PAM, the symbol rate R can be doubled with typically only a modest penalty on minimum SNR required. 
     A QAM signal is depicted as follows:
 
 s ( t )= x ( t ) cos 2π  f   c   t+y ( t ) sin 2π  f   c   t    (3)
 
     To recover the baseband x(t) signal, a digital receiver relies on coherent demodulation by multiplying s(t) by cos 2π f c t, resulting in the following: 
     
       
         
           
             
               
                 
                   
                     
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     Then a low pass filter is used to eliminate all but the baseband signal x(t). To recover y(t), s(t) is multiplied by sin 2π f c t. A low pass filter then recovers y(t). Thus QAM provides the benefit of doubling the data rate, but the required coherent demodulation increases receiver complexity. 
     Single Sideband (SSB) is an alternative QAM. In SSB, instead of transmitting two redundant sidebands of the modulated PAM signal, one of the sidebands is suppressed at the transmitter without losing the transmitted information. SSB, however, requires more complexity at the receiver to recover the baseband information from the remaining sideband. SSB also requires coherent demodulation. 
     A common method of generating SSB is by using a quadrature phase shifting Hilbert Transformer as shown in  FIG. 13 . In general, it is not possible to build one network, which can produce a 90 degree phase shift across a whole band of frequencies. However one can design two networks, a P-network and a Q-network such that the relative phase at the output of the two networks are approximately 90 degree separated across the required band. (See  IRE Transactions on Circuit Theory , June 1960, pp. 128-136, Normalized Design of 90° Phase Difference Networks, S. D. Bedrosian). 
     The exemplary quadrature phase shifter  1300  receives a baseband signal and routs the signal simultaneously to a P-network  1305  and a Q-network  1310 , the outputs of the two networks  1305 ,  1310  being about 90° out of phase with respect to each other. The outputs are each input into a respective multiplier  1315 ,  1320 , each multiplier  1315 - 1320  mixing the respective input signal with a respective carrier signal. The mixed outputs of the two multipliers  1315 ,  1320  are combined in a summer  1325  producing a SSB output signal. 
     As one approaches DC (the lower end of the band) or the upper end of the band, the error increases, which is another reason to minimize/eliminate the energy near the band edges. 
     In order to operate in the 60 GHz range, a modem must have the ability to generate a 60 GHz carrier signal. Conventional oscillator circuits and crystals cannot generate such a high frequency directly. Typically, a dielectric resonator oscillator (DRO) field-effect transistor (FET) is used to generate a 7.5 GHz signal that is then multiplied by a factor of eight (8) to achieve a 60 GHz carrier signal. Unfortunately, an affordable DRO FET has an accuracy deviation of approximately 0.1% and medium jitter of approximately 5%. When the DRO FET output signal is frequency multiplied by 8 to achieve the desired 60 GHz carrier, the resulting jitter increases by the same factor of 8 to 40%, resulting in an unacceptable amount of phase noise. In another embodiment, a crystal operator operating in the sub-Gigahertz frequency band is directly multiplied up to 60 GHz. This allows for high frequency accuracy but does not solve the phase noise problem. 
     Jitter essentially causes sine to cosine and cosine to sine conversions in a signal that result in closure of the “eye” diagram. If QAM is used, the resulting data point constellation is significantly distorted by the jitter, reducing the allowable error margin between data points. Thus, the SNR in such a modem must be high to compensate for the “eye” closure. With 40% jitter, the necessary SNR may be impracticable for a 60 GHz modem that has to go a reasonable distance. 
     The most expensive component of a modem is typically the power amplifier. At 60 GHz, a reasonably affordable power amplifier produces 1 dB compression at approximately 19 dBm power. Gallium Nitride (GaN) power amplifiers may produce higher power outputs before the 1 dB compression point is reached, however, the cost of making GaN wafers by epitaxy further increases the power amplifier cost. Because conventional 60 GHz wireless modems typically use QAM that is susceptible to signal compression, the output power of these modems has to be effectively reduced by 6-9 dB to stay in the linear range. This is a big penalty and the system can only operate at a 10 dBm power level while paying for a 19 dBm amplifier. 
     Other requirements on the modem were outlined in the previous section. QAM modems fail most of these requirements. 
     In a particular embodiment, the foregoing problems associated with current 60 GHz wireless modems are resolved or mitigated using various techniques described as follows. To reduce ISI, binary data bits are initially filtered into approximate Square-Root Nyquist (SRN) pulses with a 20% EB as shown in  FIG. 12 . Nyquist pulse shaping is achieved by using matched filtering with a respective SRN filter at each of the WAP  210  and STA  215 . 
     To eliminate the 9 dBm back-off caused by using QAM, Dual Rail Bipolar (DRB) coding is used instead to create the baseband signal. With DRB, a second Nyquist pulse train is added to data channel as illustrated in  FIG. 14 , effectively doubling the channel data rate. As shown, two dual rail signals  1400 ,  1410  are multiplexed together forming a composite DRV waveform. The compression problem associated with QAM is also nearly eliminated using DRB with Automatic Gain Control (AGC) to mitigate non-linear distortion of the baseband signal. the As shown in  FIG. 15 , shows the spectral contributions of the individual components of the components of the baseband signal. Namely, a Nyquist pulse spectrum  1500  and a coding spectrum  1510  are overlayed. When combined, the resulting spectrum of the encoded signal is illustrated in  FIG. 16 . The resulting bandwidth has been reduced and substantially contains most of the energy 
     For a data rate of 1 Gbps, using 20% excess Bandwidth Nyquist Pulses, the resulting bandwidth is approximately 600 MHz. Further, as the signal dwindles to zero power at DC and at 500 MHz, the 20 dB down signal occupancy  1600  which contains almost all the signal power is confined to approximately 20-480 MHz. Thus the modem has packed 1 Gbps into 460 MHz of bandwidth resulting in a bit density of about 2.2 bps/Hz, which is better than QPSK (4 QAM) and straight Bipolar transmission. 
     To transmit the 500 MHz coded baseband information, the signal must be moved to or used to modulate a carrier signal to an IF and, eventually, to the 60 GHz RF modem output. The resulting modulated signal, however, will effectively double the base bandwidth back to 1 GHz due to the modulation process. For example, if the DRB signal is multiplied by a 2 GHz IF signal during the modulation process, the resulting signal will have an upper sideband to 2.48 GHz (2 GHz+0.48 GHz) and a lower sideband at 1.52 GHz (2 GHz−0.48 GHz). The combination of the two approximately 500 MHz sidebands results in an approximately 1 GHz bandwidth signal. The embodiment here employs SSB (or equivalently VSB) to counter the doubling of the DRB bandwidth during modulation by eliminating (or filtering) all (or a significant portion) of one of the signal sidebands before transmission. 
     SSB (and VSB) requires coherent demodulation at the receiver to recover the baseband signal. When performing coherent demodulation, however, the receiver frequency must match the transmitted carrier frequency to prevent degradation and distortion of the signal. To enable synchronization of the carrier signals between the transmitter and receiver, the transmitter may add a pilot signal (α 1 ) at half the baud rate (500 MHz in the 1 Gbps example) to the remaining SSB or VSB sideband before transmission. The receiver then uses a notch filter to recover the pilot from the received signal that ensures coherency between the carrier at the transmitter and receiver since the pilot and the signal undergo equal phase transformations over the essentially dispersion free channel. After the receiver uses the pilot for coherent demodulation, the receiver recovers the encoded DRB signal in the time domain as shown in  FIG. 17 . A demodulator  1700  with pilot-signal recovery receives an SSB modulated signal including a pilot signal (x). The signal is directed simultaneously to a notch filter  1705  and to a multiplier  1710 . The notch filter  1705  is turned to the pilot signal frequency. An output of the notch filter including the pilot is input to the multiplier  1710 . The output of the multiplier  1710  is routed through a low pass filter  1715  providing the received baseband signal. 
     A second pilot signal (α 2 ) may also be used as shown in  FIG. 20  to enable clock and data recovery at the receiver. In this instance, the second pilot, or Pilot  2 , is located a frequency corresponding to ½ T at the baseband while the first pilot, or Pilot  1 , is located at a frequency corresponding to DC in the baseband signal.  FIG. 18  shows an embodiment of an incoming IF signal  1800  having two pilots, the recovery of the first pilot al can be accomplished using a notch filter and used in coherent demodulation of the signal. The second pilot is inserted at the transmitter at a well-defined power level relative to the signal. The second pilot α 2  is recovered and its power measured. This information can be used to drive an AGC for the received signal. A demodulator  1900  with dual-pilot signal recovery receives an SSB signal including the two pilots α 1 , α 2  ( FIG. 18 ). The received signal is routed to a wideband filter  1925  adapted to filter the second pilot α 2  and to one input of a first mixer  1910 . The received signal is also coupled to a second input of the first mixer  1910 , the output of the mixer being filter  1915  and routed to a second stage including a notch filter  1920 , second multiplier  1925  and low pass filter, similar to the demodulator of  FIG. 19 . 
     A slicing level used in detection of the data can also be derived from the received power level of the second pilot signal. The receive SRN filter eliminates the second pilot from the AGC&#39;d signal. Detectors typically include a slicer determining received data according to an energy level for example, a slicing level can be derived from the pilot signal of  FIG. 20 . A Group delay equalizer equalizes the combined group delay of the transmit SRN, The Hilbert Transformer, and the receive SRN. The slicer then slices the two rails using two dual-comparators banks to recover the two rails. One of the dual comparators slice the positive going data bits of a rail while the second comparator slices the negative going data bits of the rail. One bank&#39;s positive and negative outputs are latched on the positive going edges of the recovered half Baud rate clock (Pilot  2  based) and the other bank&#39;s positive and negative outputs are latched on the negative going edges of the same clock. This recovers the two rails of the DRB signal. 
     One type of notch filter that may be used is based on a ceramic coaxial resonator that has a high Q, defined as the center frequency of the filter divided by the bandwidth between the −3 dB points. The Q of a coaxial resonator is typically 300 to 500 while the center frequency is set by the filter length where L=λ/4 and λ is the wavelength of the center frequency to be filtered. While a notch filter passes all frequencies except those in the stop band centered on the center frequency as shown in  FIG. 21 , the output signal magnitude is maximized at the center frequency as shown in  FIG. 22  because the notch filter is effectively in parallel with the output load. 
     Unfortunately, due to the high Q value, the output of the notch filter may be substantially affected by only small frequency errors (relative to the filter&#39;s center frequency) in the incoming signal. The phase response of such a high Q synchronously tuned band pass filter has a very steep slope at the center frequency. Tens of degrees of phase error can be created by a few ppm errors in frequency. This can cause significant problems with coherent demodulation. A novel approach to eliminating this problem is to place two notch filters in series with the center frequency of each filter offset by a respective frequency difference (Δ) above and below the targeted center frequency. The slope of the phase response at the center frequency of such a filter can be considerably reduced relative to the synchronously tuned filter by the right choice of Δ. For example, to filter a center frequency of 1 GHz, the Δ value may be set at about 50 KHz such that the first and second notch filters have center frequencies of 1.005 GHz and 0.995 GHz respectively, e.g., 100 KHz bandwidth, as shown in  FIG. 23 . The combination of notch filters effectively lowers the overall phase error across the 50 ppm band to allow for 1-2° phase error. 
       FIG. 24  provides a functional block diagram of one embodiment of a wireless modem transmitter  2400  operating within the 57-64 GHz range that may be implemented within an STA  215  or WAP  210  of the wireless network embodiment described above. While the exemplar transmitter uses SSB, VSB may also be employed by replacing the SSB Hilbert Transformer with a VSB filter. The operation of the IF back end and 60 GHz front end of the transmitter and receiver are described as follows. 
     A data source  2405  provides data on two rails  2406 ,  2408  each of which have been separately Bipolar encoded, i.e., alternate 1s have alternating polarity. The positive 1s are put out on one sub-rail while the negative 1s are put out on another sub-rail. The four sub-rails couple the data source  2405  to a coupling circuit  2410  that merges them together to add the two positive 1 sub-rails together and subtract the negative sub-rails resulting in the DRB NRZ pulsed signal (“eye” diagram) as shown. This signal is then filtered by a SRN Filter  2415  and then fed to a Hilbert Transformer  2420  employing two Bedrosian networks: a P-network  2422  and an Q-network  2424  to achieve a relative 90-degree phase between the Q-rail and the P-rail. Pilot two α 2  at ½ T Hz is then added to the P-rail output. 
     The augmented P rail output and the Q rail output are then input into a quadrature up-converter  2425 , yielding at its output an SSB signal at the IF frequency. The final SSB signal is then loosely filtered in a bandpass filter  2430  to eliminate the unwanted image of the Pilot  2  frequency (f IF +½ T in this example). At the SRN filter  2415 , DC is added resulting in the formation of Pilot  1  at the IF frequency after the up-conversion. 
     While  FIG. 24  shows the case where the quadrature up-converter  2425  subtracts the sine rail from the cosine rail to preserve the lower sidelobe, one skilled in the art would recognize that by reversing the operation the upper side lobe could have been preserved instead. In this case the band pass filter  2430  would have suppressed the Pilot  2  image at frequency f IF −½ T instead. Finally, the IF is SSB filtered and SSB up-converted to 60 G Hz using an up converter  2435  with Band pass filtering which is non-intrusive to the signal band with respect to group delay (big frequency gaps are present to simplify this filtering). 
       FIG. 25  demonstrates a possible receiver  2500  for the modem. The operation has been well enough outlined above so that the function of each block should be immediately clear. A millimeter wave signal (e.g. at 50 GHz) is received and down converted to an IF at a down converter  2505 . The IF signal is routed to a high-Q filter  2510  and to one input of a multiplier  2515 . The output of the filter provides the first pilot signal and is routed to the second input of the multiplier  2515 , forming a coherent demodulator. The demodulated signal is routed through a second filter  2520  to filter out the second pilot signal and to an AGG unit  2530 . The output of the second filter  2520  is routed to a power detector  2525  and to a phase adjust circuit  2550 . One output of the power detector  2525  controls the AGC unit  2530 . Two other outputs are routed to a slicer  2545 . The output of the phase adjust  2550  clocks data into latches provided within the slicer  2545 . Dual-rail bipolar data is available at the output of the slicer  2545 . The final output of the slicer  2545  regenerates the four wire digital interface at the data source at the data sink and the modem operation is complete. Using standard communication theory, it can be shown at this modem will operate at approximately 10 −6  error rate at about 13 dB SNR. 
       FIGS. 26A through 26D  illustrate an exemplary realization of the SRN filter  2415 , the Hilbert Transformer  2420 . The SRN filter  2415  consists of the circuit shown in  FIG. 26A  coupled to the circuit shown in  FIG. 26B . This second circuit is used to correct the sin(x)/x dispersion present in the NRZ pulses put out by the coupling circuit  2410 . The SRN filter includes a network of five capacitors C 1 -C 5  and two inductors L 1 , L 2 . The circuit  2600  includes an inverting amplifier receiving an input with the amplifier output coupled through a variable potentiometer, a variable inductor L 1  and a variable capacitor C 1 . 
     The GD equalizer of  FIG. 26C  includes two circuit branches. A first branch receives an input in a mesh circuit including an inductor L 1  and two capacitors C 1 , C 2 . The first mesh circuit is coupled between the two capacitors to ground through a third capacitor C 3 , and two inductors L 2 , L 3 . A second branch is coupled to the first at one end of the mesh circuit and a second mesh circuit including a transformer M 1 ′ and a capacitors C 1 ′. The second mesh circuit is coupled between the transformer to ground through a second capacitor C 2 ′, and two inductors L 3 ′, L 4 ′. 
     The Hilbert transformer includes a quadrature up-converter receiving balanced inputs, each input including a network of series transformers each shunted to ground through a respective capacitor. 
       FIG. 27  shows an exemplary slicer output in response to a single 1 put into the transmitter. From the 1 ns periodic zero crossings around the peak value it should be clear that the resulting “eye” diagram will exhibit a very open eye and the modem will perform close to theory. 
     It should be obvious to one of ordinary skill in the art how to construct, implement, integrate, and manufacture the various components of the wireless modem embodiments such as the filters, multipliers, adders, amplifiers, pulse shapers, and other components. Also, the features of the embodiments of the modem apparatus described herein may apply to any bandwidth limited channel operating at a high frequency with a high data rate. 
     Antenna Design and Configuration 
     The wireless system employs two distinct antenna types. At the Wireless Access Point (WAP)  210 , a cosecant squared (csc 2 ) antenna is used. This provides 90° azimuth coverage and cosecant-squared elevation coverage. This ensures that the signal level anywhere within the 90° sector remains virtually constant. At the station (STA)  215 , an 8-sectored antenna is used. This provides 360° azimuth coverage, 45° elevation coverage per antenna element. In order to minimize the reflections from objects within the operating space, circular polarization is used. 
     (A) WAP Antenna 
     The challenges for WAP antenna are:
         (i) Operating bandwidth in excess of 10%   (ii) The cosecant-squared beam shaping   (iii) A ripple free, 90° azimuth pattern   (iv) Broadband polarization operation with &gt;15 dB cross polarization isolation   (v) Low loss       

     The approach to realizing the WAP antenna utilizes a pillbox style antenna with a polarizer placed at the aperture. A pillbox antenna is an H-plane offset-feed architecture, which avoids aperture blockage and, thus, allows good shaping (csc 2 ) of the elevation far-field pattern. It provides wide bandwidth, side lobe level control in elevation and avoids the aperture phase error that arises in conventional sector horns as the aperture length increases. 
     The azimuth pattern is controlled by placing choke slots along the aperture and the polarization of the antenna is changed from linear to circular by using a dielectric structure placed in front of the aperture. The circular polarization (CP) can be right hand (RHCP) or left hand (LHCP) depending on the orientation of the dielectric structure. 
     Elevation Reflector 
     An exemplary WAP antenna  3400  includes a cosecant-squared reflector  3410  fed from an offset feed  3405 . An exemplary reflector surface  3410  is shown in  FIG. 28A . A normalized patter of the elevation gain is shown in  FIG. 28B  for the exemplary antenna. The efficiency of the csc 2  pattern is about 50% (M. Skolnick,  Introduction to Radar Systems , McGraw Hill, New York, 1980, p. 259) and it is given by eff=2−(θ 1 /tan θ 2 ), θ 1  and θ 2  are the reflector angle where the csc 2  shape starts and stops respectively. Normally θ 1  and θ 2  are between [2, 5] and [20, 30] degrees respectively. Note that the efficiency decreases as θ 2  gets larger. 
     The shaped reflector profile is synthesized by using the next two transcendental equations (R. S. Elliot, Antenna Theory and Design, Prentice Hall, Englewood Cliffs, 1981, pp. 298-504). Setting up a table of θ values, equation (5) gives φ in the range [φ 1 , φ 2 ]. Now, using equation (6) a table of ρ(φ) values is obtained which is the reflector profile. ρ(φ) is then converted to Cartesian coordinates and used to model the antenna. 
     
       
         
           
             
               
                 
                   
                     
                       
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     The far-field pattern is determined applying the aperture field method (Elliot, ibid, pp. 508-518) at a reference plane in front of the aperture. The total radiated field, this is given by equation (7), which takes into account the feed back radiation 
     
       
         
           
             
               
                 
                   
                     
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     D feed  and D ref1  are the peak gains of the reflector and feed. See Holzman (Eric L. Holzman,  Pillbox Antenna Design for Millimeter - Wave Base - Station Applications , IEEE AP Magazine, Vol. 25, No. 1, February 2003, pp. 27-37) for the rest of the definitions. For the reflector case, the directivity is given by:
 
 D   ref1 =4π[ y (φ 1 )− y (φ 2 )](0.866λ 0 /θ 3 dB )/λ 0   2    (8)
 
     The reflector profile, equations (5) and (6), and the far field pattern equation (7) are determined with a Mathcad routine and the reflector along with the aperture is modeled with a full 3D EM software, CST Microwave Studio. 
     Azimuth Aperture Design 
     To reduce the strong edge diffraction along the radiating aperture and to get and acceptable input match, the aperture is flared and ground planes are placed on either side of the aperture. The azimuth pattern beamwidth can be modified by properly placing choke slots, ˜λ 0 /4 deep, parallel to the main aperture. The combination of the diffracted fields at the edges and the main aperture field creates a pattern that is broader than the main aperture and at the same time shape the skirt of the pattern. 
     Linear-to-Circular Polarizer Design 
     A linear-to-circular polarization converter can be achieved by different structures; the most common ones are discussed in Johnson (Richard C. Johnson,  Antenna Engineering Handbook , Third Edition, McGraw Hill, New York, 1993, pp. 23-25 to 23-28). Realizing these structures at millimeter wave frequencies become challenging, besides the dimensions getting too small, they require very tight tolerances or anisotropic materials which increases the mechanical complexity and the manufacturing cost. 
     When a linearly-polarized field propagates through the planar structure  3500  in  FIG. 29A  at a 45° angle relative to the slots, the linear field is decomposed into two orthogonal components, parallel (E ∥ ) and perpendicular (E ⊥ ) as shown in  FIG. 29B . 
     By properly selecting the dimensions of the slots, as shown in  FIG. 3A , the E 195   component is delayed φ ⊥ degrees and the E ∥  component is advanced φ ∥  degrees. In order to get a circularly polarized field the total field difference between the two components must be 90 degrees as shown in  FIG. 29C . Using a low dielectric constant material, such as a cross-linked polystyrene microwave plastic (i.e., REXOLITE®, having a relative dielectric constant of about ε r r=2.548 available from C-Lec Plastics, Inc. of Philadelphia, Pa.), and the structure in  FIG. 35A , it is not possible to achieve a 90° phase difference, to do so, it will require a high dielectric constant material (er˜9), which will make the all dimensions smaller, complicating the fabrication of such a device at higher frequencies, such as millimeter-wave applications. 
     To keep all the dimensions relatively large, so that the manufacturability becomes easier, REXOLITE® dielectric material is used. In order or achieve the 90° phase difference, slots are placed on both sides of a solid piece of dielectric material  3600  whose thickness is λ r , as shown in  FIG. 30A . The polarizer is designed assuming a far field plane wave but to keep assembly and manufacturing simplicity it is placed ˜λo away from the aperture, which is within the near field range. In simulations and measurements an AR ripple is noticed, this may be due to unaccounted phase error correction due to the proximity of the polarizer to the aperture. 
     Simulation Results 
     An exemplary antenna  2400  was designed with the aid of a simulation program, such as CST Microwave available from Computer Simulation Technology of Wellesley Hills, Mass. The reflector  3410  and feed  3405  were first designed as a linearly polarized antenna. The pattern was optimized for beam shape and pattern, particularly in the vertical plane. The polarizer  3600  was designed as a separate element. Due to the size and complexity of the combined linear antenna  2400  and polarizer  2600  only simulations on a short section of the aperture and polarizer was simulated. Simulation results of the linear far-field patterns are provided in  FIGS. 31A-C  at frequencies 57, 61 and 64 GHz respectively. 
     
       
         
           
               
               
               
               
             
               
                 TABLE 1 
               
               
                   
               
               
                 Frequency GHz 
                 Peak Gain (dB) 
                 EL HPBW (deg) 
                 AZ HPBW (deg) 
               
               
                   
               
             
            
               
                 57 
                 22.5 
                 2.1 
                 79 
               
               
                 61 
                 22.7 
                 2.1 
                 75 
               
               
                 64 
                 22.9 
                 2.1 
                 73 
               
               
                   
               
            
           
         
       
     
     A summary of the simulated patterns is provided in Table 1. Elevation performance simulation results are illustrated in  FIG. 31D  for the same frequencies. 
     The simulation results of the polarizer for a partial aperture are shown in  FIGS. 32A-32C  for 57, 61 and 64 GHz. 
     The challenge with a circularly polarized cosec squared antenna is further complicated when the azimuthal beamwidth is in excess of 45 degrees. In the case of the WAP antenna  3400  the 90 degree beamwidth and the operating bandwidth required some compromise. The result however over the bandwidth is good. Two cases are shown in  FIGS. 33A and 33  B, one with choking grooves  FIG. 33B , which provide improved beamwidth performance and another without choking grooves  FIG. 33A , which limit the beamwidth in azimuth. As the groove can later be removed the configuration with grooves was chosen for the first set of measurements 
     Measured Results 
     Construction 
     A test antenna and polarizer was built as shown in  FIG. 34 . The reflector structure was made as two split blocks containing the feed horn, feed waveguide and reflector surface. This is extremely integrated and lends itself to high-volume, low-cost manufacture using metallized injection molded parts. The polarizer was made from a single piece of machined plastic—again readily capable of being injection molded. Antenna performance was measured using a 57 to 64 GHz, 1 kHz modulated source. Measurement results of the Azimuth and elevation performance are provided in  FIGS. 35A and 35B  respectively. 
     Using the outlined procedure, a 60 GHz pillbox antenna and a linear-to-circular polarization converter were designed. The antenna has a csc 2  shape in elevation and an azimuth beamwidth of 90° with a peak linear gain of 23 dBi. The polarizer has about 10% BW for an AR≦3 dB over an angular range of ±45′, the best AR is presented at the center of the band and slowly degrading at the edges of the band. Easy manufacturability is another main characteristic of the polarizer, which is very attractive at millimeter-wave frequencies. Predicted and measured data has excellent agreement. 
     Cross Shaped Horn 
     A cross-shaped horn  3800  is illustrated in  FIGS. 36A and 36B . The horn  3800  has a square waveguide input  3805  and a “cross” aperture shape  3810  as shown in  FIG. 1 . The aperture field distribution is approximately equal to the dominant TE mode of a cylindrical waveguide (N. Toyama,  A cross - shaped horn and a square waveguide polarizer for a circularly polarized shaped beam antenna for a broadcasting satellite , IEEE MTT-S International Microwave Symposium, Volume 80, Issue 1, Mary 1980, pp. 299-301). 
     In order for the antenna to generate a low aspect ratio (AR), without using any phase error correction mechanism, the aperture  3810  should be symmetric on both E-plane and H-plane. To obtain a minimum AR level, a ratio (N. Toyama,  Symmetrical crossed horn for a circularly polarized multi-beam reflector antenna , IEEE AP Transactions, Vol. AP 31, No. 1, January 1983) of the aperture dimensions ‘a’ and ‘b’ can be optimized. An optimized ratio was found to be:
 
 b/a= 0.18   (9)
 
     Selecting a=b, the aperture becomes square and the E-plane beam width is smaller than the H-plane beam width. On the other hand, by selecting a&lt;&lt;b the opposite occurs and symmetry is not achieved. The flare length F must larger than b. 
     If a standard rectangular waveguide aperture is used as the main radiating aperture, not only would it be bigger than the cross-shaped horn, but a phase correction would be required. The phase correction would be done by using a dielectric lens or some sort of fins configuration. By taking this approach, not only would the whole antenna get bigger but also the assembly would be more involve. 
     Septum Polarizer 
     A septum polarizer  3900  is a four-port device, as shown in  FIG. 37A . The device  3900  includes a rectangular housing  3902  with a linear septum  3904  at one end and extending partially into the housing. Two rectangular waveguide ports  3905 ,  3910  (referred to a ports ‘L’ and ‘R’) are defined at one end being separated by a septum  3904 . A square aperture  3915  is defined at the other end of the device, which supports two orthogonal modes (H. Schrank,  The septum polarizer , IEEE APS Newsletter, October 1983, pp. 23-24), each mode constitutes one port. Any linearly polarized signal fed into any of the two rectangular ports is equally split into two orthogonal components and the septum length introduces a 90° phase delay on one of these components, creating in this way a circularly-polarized (CP) field in the square waveguide section. The CP field can be RHCP or LHCP if the signal is fed into the port R or L respectively. This mechanism is illustrated in  FIG. 37B and 37C  and described as follows. 
     Assuming the square aperture is excited with a field parallel to the septum  3904 , the septum  3904  transforms the field into two odd-mode fields. If the square aperture  3915  is excited with a field perpendicular to the septum  3904 , the field is transformed into two even-mode components. When the two modes, parallel and perpendicular, simultaneously exist at the square aperture, cancellation at one of the rectangular (i.e., L and R) ports  3905 ,  3910  occurs only if the amplitudes are the same and the phase difference is either 0° or 180°. 
     If a RHCP field excites the square aperture  3915 , the vertical component will be 90° phase delayed by the septum  3904  relative to the horizontal component, resulting in field cancellation at port L  3905  and adding in phase at port R  3910 . The septum polarizer  3904  works as a polarization diplexer if both RHCP and LHCP are present at the square aperture  3915 . A rigorous analysis of the septum polarizer in found in Bornemann (J. Bornemann,  Ridge waveguide polarizer with finite and stepped - thickness septum , IEEE MTT Transactions, Vol. 43, No. 8, August 1995). 
       FIGS. 40A and 40B  show an antenna configuration  4000  including the cross-shaped horn  3800  combined with the septum polarizer  3900 . As illustrated, the septum polarizer  3900  is coupled to the feed point of the horn  3800 . As illustrated, the rectangular port  3915  is coupled to the horn  3800 . 
     Simulation Results 
     A simulation was run for a right-hand circularly polarized, cross-shaped, square horn design using CST Microwave. The simulation results show a directive gain of about 17 dBi, and an Axial Ratio of 2 maximum across the band of 55-66 GHz. This configuration of antenna has very good broadband performance when compared to a patch arrangement. Unfortunately it is more complicated to construct. It is believed that a hybrid can be used which will have the best attributes of both. 
     Measured Results 
     Measured results for a right-hand CB and left-hand CB square cross-shaped horn are provided in  FIGS. 39A and 39B  respectively show very good agreement with the simulated data. The axial ratio shows some degradation but this may be due partly to the resolution and accuracy of the axial ratio measurement set up. 
     Measured gain and aspect ration for the antenna of  FIGS. 40A and 40B  are provided in  FIGS. 41A and 41C  at frequency of 57, 60 and 64 GHz respectively 57, 60, and 64 GHz 
     Next Generation of Station Antenna 
     It is believed that a hybrid (waveguide and printed patch) antenna configuration  4100  for the Station side of the link will prove to be the best combination of good gain and axial ratio performance and low cost construction as shown in  FIG. 42 . The concept is to feed a waveguide crossed horn  4105  with circularly polarized patch antenna  4110  placed at the throat of the horn  4105 . The horn  4105  can be very inexpensively constructed using a metallized injection molded part. 
     A suspended circularly polarized patch antenna with an inverted microstrip feed is illustrated in  FIG. 43A . The patch can be configures as a circle, as shown in  FIG. 43B  or as a square, as shown in  FIG. 43C . The microstrip feed line is located on one side of a dielectric substrate and the patch element located on the opposite side. The dielectric can be a duroid having a relative dielectric constant of about 2.2. Such a design offers lower conductor loss, while also providing increased antenna efficiency. Fabrication is also simplified as a single substrate layer is used for both the patch and the feed line. Some complexity is involved in interfacing the antenna with conventional circuitry and it requires a low-loss transition from a conventional to inverted microstrip line. An exemplary embodiment of a circular patch is illustrated in  FIGS. 44A and 44B . The resulting return loss is illustrated in  FIG. 44C .  FIGS. 45A-45H  illustrate exemplary performance parameters of the device of  FIGS. 44A and 44B . 
       FIG. 46B  shows an exemplary transition from a standard microstrip to inverted microstrip. Preferably, the transition is non-contacting with an insertion loss of about 0.9 dB and having an acceptable standing-wave ratio bandwidth.  FIG. 46A  shows exemplary return loss of the device of  FIG. 46B . 
       FIGS. 47A and 47B  show an exemplary suspended patch antenna with inverted microstrip feed combined with a cross-shaped, square horn antenna. As shown, a patch antenna is placed at the narrow end of the horn antenna, with the patch facing into the cavity of the horn.  FIGS. 48A through 48F  illustrate exemplary performance parameters of the device of  FIGS. 47A and 47B . 
     In order to support the square or rectangular cell wireless network architecture described earlier while also supporting transmissions at mmwLAN frequencies, certain wireless antenna configuration and design embodiments are described as follows. 
     To enable the square cell architecture of the wireless network embodiment, each WAP transmitter (Tx) and receiver (Rx) is arranged according to  FIG. 2A . Because each WAP  210  is located within a corner of each square cell  200 , each WAP  210  has two interface sectors with each sector containing a Tx and Rx. The total coverage of each WAP  210  is greater than 45 degrees. In some embodiments the WAP coverage is about 90 degrees. Thus, with a WAP  210  in each corner of a square cell, the complete area of the cell may be covered by all four WAPs  210 . The WAP Tx and receiver Rx antennas are typically bistatic with right hand polarization. 
     Each WAP antenna may use a cosecant-squared beam shape with an elevation greater than or equal to 1.5 meters vertical distance between the WAP and an STA as shown in  FIG. 49 . The cosecant-squared beam shape has the advantage of maximizing the transmission range, minimizing wasted energy, and minimizing RF exposure, especially in the mmwLAN range of frequencies. The beam shape  2910  generated, as also illustrated, possesses a θ 3 dB  AZ=22.5° per sector and θ 3 dB  EZ=12° per sector with main lobe 14° above horizontal (see  FIG. 49B ). The WAP antenna peak gain is typically 23.5 dBi per sector. The WAP antenna may be implemented as a horn antenna, antenna array, or on a printed circuit. While printed circuit antennae may be lossy, they are easy to interface with mmwLAN circuits. 
     To interface with a WAP  210  of the wireless network embodiment, each STA modem may use a four sector antenna  5000  of bistatic design as shown in  FIG. 50  to provide the STA with 360° coverage. Each STA  215  may use a cosecant-squared beam shape with a θ 3 dB  AZ=90° per sector, 360° per antenna and θ 3 dB  EZ=12° per sector with main lobe 14° above horizontal. The antenna  5000  may be realized as a horn or printed circuit. As described previously, the STA typically searches for all color-coded WAPs and sectors, measures the strength of the signal emanating from each of the four color-coded WAP within a cell, and then selects the two strongest signals for communication with the network. 
     In certain circumstances, to counter excessive attenuation caused by an obstacle such as a wall, a wireless Network Extender (Nex)  5100 , as illustrated in  FIG. 51 , may be employed to boost the signal strength and extend the range of a WAP. A Nex  5100  is a physical layer device or set of devices wherein horn antennas are used to penetrate an object such a wall  5110  and then retransmit the original signal on the exiting side of wall. Functionally, each Nex  5100  provides signal repeating, much like an Ethernet repeater, in both directions to and from a WAP  210 . To improve signal penetration through an object, the Nex  5100  may utilize one or more lenses  5120  within each spot focusing antenna. Also, one or more amplifier circuits  5125  may be used to boost the received signal from a horn antenna before delivery of the amplified signal to a spot focusing antenna  5130 . Two pairs of spot focusing antenna  5130 , horn antenna  5105 , and amplifier units  5125  may be employed to provide signal amplification and repeating in both directions across an obstacle. 
       FIG. 52  provides an illustrative example comparing the range extension provided by the WAP extender  5100  to counter the rapid attenuation of signals in the mmwLAN range of frequencies. A signal propagates from a WAP  210  to a STA  215  through three walls  5200   a ,  5200   b ,  5200   c . Two of the walls provide about 10 dB of attenuation and the third wall provides about 20 dB. The WAP antenna has a gain of about 24 dBi and the STA antenna has a gain of about 15 dBi. Without the extender  5100 , the signal travels for a range of about 40 meters with a 0 dB margin. With the extender, however, the signal travels for a range of about 180 meters, while still retaining an additional 10 dB of margin. 
     It should be obvious to one of ordinary skill in the art how to construct, implement, integrate, and manufacture the various components of the wireless modem antenna components described herein. 
     With the improvements within the modem embodiments, 23 dBm is realized as opposed to 10 dBm in earlier mmwLAN modems. Using the novel cosecant-squared antenna design, a 41 dB gain is realized at the antenna as opposed to 18 dB in prior unsophisticated antennas. Furthermore, frequency, space, and time diversity provided by the network architecture further improve SNR by approximately 9 dB. Taking the improvement from each of the embodiments as a whole, a wireless communications system is realized with 48 dB gain, possibly enabling up to 600 meters of coverage in the mmwLAN frequency range as shown in  FIG. 4 . 
     While this invention has been particularly shown and described with references to example embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the scope of the invention encompassed by the appended claims.