Patent Publication Number: US-2023163731-A1

Title: High frequency amplifier

Description:
TECHNICAL FIELD 
     The present disclosure relates to a high frequency amplifier. This application claims the benefit of priority from Japanese Patent Application No. 2020-076598, filed on Apr. 23, 2020, the entire contents of which are incorporated herein by reference. 
     BACKGROUND ART 
     Regarding a high frequency amplifier, for example, Patent Literature 1 discloses a technology related to a field effect transistor (FET). This field effect transistor includes a plurality of amplifier elements for amplifying a radio frequency (RF) signal, and a matching circuit connected to input ends of the amplifier elements and an input terminal of a package via bonding wires and performing impedance conversion. 
     CITATION LIST 
     Patent Literature 
     [Patent Literature 1] Japanese Unexamined Patent Publication No. S63-86904 
     SUMMARY OF INVENTION 
     The present disclosure provides a high frequency amplifier. This high frequency amplifier includes a first transistor, a second transistor arranged side by side with the first transistor in a first direction, and a third transistor arranged side by side with the second transistor in the first direction on a side opposite to the first transistor; a first drain pad electrically connected to a drain electrode of the first transistor, a second drain pad electrically connected to a drain electrode of the second transistor, and a third drain pad electrically connected to a drain electrode of the third transistor; a matching circuit pattern including a first transmission line electrically connected to the first drain pad, a second transmission line electrically connected to the second drain pad, and a third transmission line electrically connected to the third drain pad, the matching circuit pattern performing impedance matching of a radio frequency signal with respect to each of the first transistor, the second transistor, and the third transistor; a first wire that electrically connects the first transmission line and the first drain pad to each other, a second wire that electrically connects the second transmission line and the second drain pad to each other, and a third wire that electrically connects the third transmission line and the third drain pad to each other; and a wiring pattern electrically connected to the first drain pad via the first transmission line and the first wire and electrically connected to the second drain pad via the second transmission line and the second wire. An effective impedance of the second wire is larger than an effective impedance of the first wire. The matching circuit pattern has an asymmetrical external shape with respect to a second virtual straight line orthogonal to a first virtual straight line connecting a first connection point having the first wire connected thereto in the first transmission line and a second connection point having the second wire connected thereto in the second transmission line to each other and passing through a median point between the first connection point and the second connection point. An electrical length of the second transmission line is shorter than an electrical length of the first transmission line. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG.  1    is a plan view illustrating an internal constitution of a high frequency amplifier  1 A according to a first embodiment and a high frequency amplifier  1 B according to a second embodiment. 
         FIG.  2    is a plan view illustrating an amplifier element  11  and a matching circuit  50 . 
         FIG.  3    is an enlarged plan view of one matching circuit pattern  52  in  FIG.  2   . 
         FIG.  4    is an explanatory perspective view of each electrical length of bonding wires W 41 , W 42 , W 43 , W 44 , W 45 , W 46 , W 47 , and W 48 . 
         FIG.  5    is a Smith chart showing each impedance of the bonding wires W 41 , W 42 , W 43 , W 44 , W 45 , W 46 , W 47 , and W 48 . 
         FIG.  6    is a graph showing an imaginary portion of each impedance of the bonding wires W 41 , W 42 , W 43 , W 44 , W 45 , W 46 , W 47 , and W 48 . 
         FIG.  7    is an explanatory view of a relationship between a length of a physical path through which electromagnetic waves pass and a phase. 
         FIG.  8    is an explanatory view of a relationship between a length of a physical path through which electromagnetic waves pass and a phase. 
         FIGS.  9 A and  9 B  are an explanatory view of a distribution state of currents in microwaves. 
         FIG.  10    is a view of a simulation illustrating a concentration state of currents in microwaves. 
         FIG.  11    is a view of a simulation illustrating a distribution state of electric field intensities in microwaves. 
         FIG.  12    is a plan view illustrating one reference example of a pattern  80 A through which electromagnetic waves pass. 
         FIG.  13    is a plan view illustrating another reference example of a pattern  80 B through which electromagnetic waves pass. 
         FIG.  14    is a graph showing a phase difference occurring between two electromagnetic waves which have respectively passed through paths DA and DB in the patterns  80 A and  80 B. 
         FIG.  15    is a plan view illustrating a matching circuit  60 . 
         FIG.  16    is an enlarged plan view of one matching circuit pattern  62  in  FIG.  15   . 
         FIG.  17    is a Smith chart showing each load impedance in transistors  13 A,  13 B,  13 C, and  13 D of the high frequency amplifier  1 B according to the second embodiment. 
         FIG.  18    is a view showing a relationship between variation in phase of a load impedance and electric power efficiency according to the second embodiment. 
         FIG.  19    is a plan view illustrating a matching circuit  60 X of a high frequency amplifier according to a comparative example. 
         FIG.  20    is a Smith chart showing each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D of the high frequency amplifier according to the comparative example. 
         FIG.  21    is a view illustrating a relationship between variation in phase of a load impedance and electric power efficiency according to the comparative example. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Problem to Be Solved By Present Disclosure 
     In the high frequency amplifier disclosed in Patent Literature 1, a plurality of transistors are disposed in parallel in an amplifier element, and each of the transistors is connected to a matching circuit through a wire such as a bonding wire. In such a case, mutual inductance components of a plurality of respective wires receive an influence from a different wire close thereto. Accordingly, variation may occur in an effective impedance between a plurality of wires in accordance with the degree of an influence received from a different wire. At this time, since the phase of an RF signal varies between a plurality of wires, there is concern that variation may also occur in the phase of the RF signal between a plurality of transistors. In order to improve electric power efficiency of the high frequency amplifier, it is desirable to reduce this variation in phase of the RF signal. 
     Effects of Present Disclosure 
     According to a high frequency amplifier of an embodiment of the present disclosure, it is possible to reduce variation in phase of an RF signal between a plurality of transistors. 
     Description of Embodiment of Present Disclosure 
     First, details of an embodiment of the present disclosure will be enumerated and described. A high frequency amplifier according to the embodiment includes a first transistor, a second transistor arranged side by side with the first transistor in a first direction, and a third transistor arranged side by side with the second transistor in the first direction on a side opposite to the first transistor; a first drain pad electrically connected to a drain electrode of the first transistor, a second drain pad electrically connected to a drain electrode of the second transistor, and a third drain pad electrically connected to a drain electrode of the third transistor; a matching circuit pattern including a first transmission line electrically connected to the first drain pad, a second transmission line electrically connected to the second drain pad, and a third transmission line electrically connected to the third drain pad, the matching circuit pattern performing impedance matching of a radio frequency signal with respect to each of the first transistor, the second transistor, and the third transistor; a first wire that electrically connects the first transmission line and the first drain pad to each other, a second wire that electrically connects the second transmission line and the second drain pad to each other, and a third wire that electrically connects the third transmission line and the third drain pad to each other; and a wiring pattern electrically connected to the first drain pad via the first transmission line and the first wire and electrically connected to the second drain pad via the second transmission line and the second wire. An effective impedance of the second wire is larger than an effective impedance of the first wire. The matching circuit pattern has an asymmetrical external shape with respect to a second virtual straight line orthogonal to a first virtual straight line connecting a first connection point having the first wire connected thereto in the first transmission line and a second connection point having the second wire connected thereto in the second transmission line to each other and passing through a median point between the first connection point and the second connection point. An electrical length of the second transmission line is shorter than an electrical length of the first transmission line. 
     In this high frequency amplifier, each of the first transistor, the second transistor, and the third transistor serving as a plurality of transistors is connected to the matching circuit pattern through the first wire, the second wire, and the third wire serving as a plurality of wires. In this constitution, variation may occur in the effective impedance between the plurality of wires in accordance with the degree of an influence received from a different wire. Such variation in effective impedance between a plurality of wires is manifested as variation in electrical length between the plurality of wires. Here, in this high frequency amplifier, since the matching circuit pattern has an asymmetrical external shape, the electrical length of the second transmission line is shorter than the electrical length of the first transmission line. Accordingly, in a constitution in which the electrical length of the second wire is longer than the electrical length of the first wire, an electrical length difference between the wires can be canceled out by an electrical length difference between the transmission lines. Therefore, variation between the electrical length from the first transistor to the wiring pattern and the electrical length from the second transistor to the wiring pattern is reduced. If variation in electrical length between a plurality of wires is reduced, variation in effective impedance between the plurality of wires is reduced. Therefore, due to this constitution, variation in phase can be reduced between the first transistor and the second transistor. 
     In the foregoing high frequency amplifier, the matching circuit pattern may include a first pad including a first corner portion and a second pad including a second corner portion. The first transmission line may be constituted of a part including the first corner portion in the first pad. The second transmission line may be constituted of a part including the second corner portion in the second pad. The second corner portion may be chamfered by a chamfering amount larger than a chamfering amount of the first corner portion. According to knowledge of this inventor, currents (electric field) in a radio frequency are likely to be concentrated at an outer edge of a conductor. For this reason, in the first transmission line, a radio frequency signal is propagated along the first corner portion in the first pad. In the second transmission line, a radio frequency signal is propagated along the second corner portion in the second pad. Here, since the second corner portion is chamfered by a chamfering amount larger than the chamfering amount of the first corner portion, the path including the second corner portion becomes shorter than the path including the first corner portion, and thus a constitution in which the electrical length of the second transmission line is shorter than the electrical length of the first transmission line may be realized. According to this constitution, the electrical length from the first transistor to the wiring pattern and the electrical length from the second transistor to the wiring pattern are likely to be subjected to matching through adjustment of the chamfering amount. Therefore, variation in phase between the first transistor and the second transistor can be more reliably reduced. 
     In the foregoing high frequency amplifier, the first transmission line may be thinner than the second transmission line. In this case, since the impedance per unit length of the first transmission line can become higher than the impedance per unit length of the second transmission line, a constitution in which the electrical length of the second transmission line is shorter than the electrical length of the first transmission line may be realized. According to the degree of thinning of the first transmission line, the electrical length from the first transistor to the wiring pattern and the electrical length from the second transistor to the wiring pattern are likely to be subjected to matching. Therefore, variation in phase between the first transistor and the second transistor can be more reliably reduced. 
     In the foregoing high frequency amplifier, a plurality of transistor groups each including the first transistor, the second transistor, and the third transistor may be provided. In this case, a high frequency amplifier having a high output can be realized. 
     In the foregoing high frequency amplifier, the first wire may be disposed at an outermost part of the transistor groups, and the second wire may be disposed between the third wire and the first wire. A length of the first wire may be equivalent to a length of the second wire. An effective electrical length from the first drain pad to the wiring pattern may be substantially equivalent to an effective electrical length from the second drain pad to the wiring pattern. In this case, since the phases substantially coincide with each other between the first transistor and the second transistor, this is particularly advantageous from the viewpoint of improving the electric power efficiency of the high frequency amplifier. 
     Details of Embodiment of Present Disclosure 
     A specific example of a high frequency amplifier according to an embodiment of the present disclosure will be described below with reference to the drawings. The present disclosure is not limited to these examples. The present disclosure is indicated by the claims, and it is intended to include all changes within meanings and a range equivalent to the claims. In the following description, the same reference signs are applied to the same elements or elements having the same function, and duplicate description thereof may be omitted. In the description, the XYZ orthogonal coordinate system indicated in the drawings may be referred to. 
       FIG.  1    is a plan view illustrating an internal constitution of a high frequency amplifier  1 A according to a first embodiment and a high frequency amplifier  1 B according to a second embodiment. First, the high frequency amplifier  1 A according to the first embodiment will be described. The high frequency amplifier  1 A includes one input terminal  2 , one output terminal  3 , an amplifier element portion  10 , a branching circuit substrate  20 , a synthetic circuit board  30 , matching circuits  40 , and matching circuits  50 . In the present embodiment, as an example, the high frequency amplifier  1 A includes two of each of the matching circuits  40  and  50 . The amplifier element portion  10  includes two amplifier elements  11 . For example, an output for one amplifier element  11  is 30 W, and for example, an output of the entire amplifier element portion  10  is 60 W. The high frequency amplifier  1 A includes a package  4  accommodating the amplifier element portion  10 , the branching circuit substrate  20 , the synthetic circuit board  30 , and the matching circuits  40  and  50 , and bonding wires W 1 , W 2 , W 3 , W 4 , W 5 , and W 6 . 
     The package  4  is made of metal and is connected to a reference potential. The planar shape of the package  4  is substantially a rectangular shape. The package  4  has side walls  4   c  and  4   d  facing each other in a first direction, and end walls  4   a  and  4   b  facing each other in a second direction. The first direction and the second direction intersect each other and are orthogonal to each other in the example. In the present embodiment, the first direction is an X axis direction, and the second direction is a Y axis direction. 
     The package  4  has a flat bottom plate  4   e  having a rectangular shape. The bottom plate  4   e  extends along a plane defined in the Y axis direction and the X axis direction. The end walls  4   a  and  4   b  stand upright along a pair of sides of the bottom plate  4   e  (sides extending in the X axis direction), and the side walls  4   c  and  4   d  stand upright along another pair of sides of the bottom plate  4   e  (sides extending in the Y axis direction). The package  4  further has a lid portion (not illustrated). The lid portion seals an upper opening formed by the end walls  4   a  and  4   b  and the side walls  4   c  and  4   d.    
     The input terminal  2  is a metal wiring pattern, and a radio frequency signal is input therethrough from the outside of the high frequency amplifier  1 A. A radio frequency signal is a signal based on a multi-carrier transmission method and is realized by superimposing a plurality of signals in which frequencies of carrier signals are different from each other. For example, a frequency band of a carrier signal is 500 MHz or lower. The input terminal  2  is provided at a middle portion of the end wall  4   a  in the X axis direction and extends from the outside to the inside of the package  4 . 
     The output terminal  3  is a metal wiring pattern, and an amplified radio frequency signal is output therethrough to the outside of the high frequency amplifier  1 A. The output terminal  3  is provided at a middle portion of the end wall  4   b  in the X axis direction and extends from the inside to the outside of the package  4 . 
     The amplifier element portion  10  is disposed on the bottom plate  4   e  of the package  4  and substantially at a middle part of the package  4  in the Y axis direction. The two amplifier elements  11  in the amplifier element portion  10  are disposed side by side in the X axis direction. Each of the amplifier elements  11  includes a plurality of transistors  13  (refer to  FIG.  2   ). For example, the plurality of transistors  13  are field effect transistors (FETs) and are high electron mobility transistors (HEMT 5 ) in one example. Each of the plurality of transistors  13  has a gate electrode, a source electrode, and a drain electrode. Each of the transistors  13  amplifies an input radio frequency signal and outputs an amplified radio frequency signal. A more specific constitution of the amplifier element  11  will be described below. 
     The branching circuit substrate  20  is disposed on the bottom plate  4   e  of the package  4 . The branching circuit substrate  20  is disposed side by side with the input terminal  2  and the amplifier element portion  10  in the Y axis direction and is positioned between the input terminal  2  and the amplifier element portion  10 . The branching circuit substrate  20  has a ceramic substrate  21  and a branching circuit  22  provided on a main surface of the substrate  21 . For example, the planar shape of the substrate  21  is a rectangular shape. A long side  21   a  thereof on one side faces the input terminal  2 , and a long side  21   b  thereof on the other side faces the amplifier element portion  10  via the matching circuits  40 . A rear surface of the substrate  21  faces the bottom plate  4   e  of the package  4 . A short side  21   c  of the substrate  21  on one side is positioned in the vicinity of the side wall  4   c  of the package  4 , and a short side  21   d  of the substrate  21  on the other side is positioned in the vicinity of the side wall  4   d  of the package  4 . That is, the substrate  21  extends from an area in the vicinity of one end to an area in the vicinity of the other end of the package  4  in the X axis direction. 
     The branching circuit  22  includes a wiring pattern  23  provided on the main surface of the substrate  21 . The wiring pattern  23  is electrically connected to the input terminal  2  via the bonding wires W 1 . A radio frequency signal is input to the wiring pattern  23  from a middle portion of the substrate  21  in the X axis direction. The wiring pattern  23  has a line-symmetrical shape with respect to a centerline of the substrate  21  in the Y axis direction. The wiring pattern  23  repeats bifurcation at a connection point with respect to the bonding wires W 1  as a starting point and ultimately reaches eight metal pads  23   a.  The eight metal pads  23   a  are arrayed side by side along the long side  21   b.  Metal pads  23   a  adjacent to each other are connected to each other via a film resistor and constitute Wilkinson couplers. Accordingly, while isolation between a plurality of gate pads  14  (which will be described below) of the amplifier element portion  10  is secured, matching of an input impedance of the amplifier element portion  10  viewed from the input terminal  2  is achieved. In the diagram, only one film resistor  23   b  is representatively illustrated. The eight metal pads  23   a  are electrically connected to the matching circuits  40  via the bonding wires W 2 . 
     The matching circuits  40  are disposed on the bottom plate  4   e  of the package  4  and are disposed between the branching circuit substrate  20  and the amplifier element portion  10  in the Y axis direction. For example, each of the matching circuits  40  is a die capacitor, and has a dielectric substrate and a circuit pattern (not illustrated) provided on a main surface of the dielectric substrate. The circuit pattern has a plurality of metal pads (not illustrated). For example, the number of metal pads is the same as the number of metal pads  23   a.  The plurality of metal pads are arrayed in a row in the X axis direction. Each of the metal pads is electrically connected to the corresponding metal pad  23   a  via the bonding wires W 2  and is electrically connected to the corresponding gate pad  14  of the amplifier element portion  10  via the bonding wire W 3 . 
     In the matching circuits  40 , T-type filter circuits are constituted of inductance components by the bonding wires W 2  and the bonding wires W 3 , and capacitance of metal pads connected between a node between these inductance components and the reference potential (bottom plate  4   e ). The matching circuits  40  perform impedance conversion through these T-type filter circuits. Normally, estimated impedances inside the transistors  13  from the gate pads  14  in the amplifier element portion  10  differ from a characteristic impedance (for example, 50Ω) of a transmission line. The matching circuits  40  convert the impedances into an estimated impedance 50Ω inside the package  4  from the input terminal  2  through the T-type filter circuits. 
     The matching circuits  50  are disposed on the bottom plate  4   e  of the package  4  and are disposed between the amplifier element portion  10  and the synthetic circuit board  30  in the Y axis direction. For example, similar to the matching circuits  40 , the matching circuits  50  are parallel flat plate-type capacitor (die capacitors). As illustrated in  FIG.  2   , each of the matching circuits  50  has a dielectric substrate  51  (refer to  FIG.  2   ) and a plurality of matching circuit patterns  52  (refer to  FIG.  2   ) provided on the dielectric substrate  51 . The planar shape of the dielectric substrate  51  is a rectangular shape having the X axis direction as a longitudinal direction. For example, the thickness (here, the dimension in a Z axis direction) of the dielectric substrate  51  is approximately 200 μm. For example, a relative dielectric constant (εr) of the dielectric substrate  51  is εr=150. Each of the matching circuit patterns  52  has a plurality of metal pads  53 . Each of the metal pads  53  is electrically connected to a corresponding drain pad  15  (which will be described below) of the amplifier element portion  10  via the bonding wire W 4  (wire) and is electrically connected to a corresponding metal pad  33   a  (which will be described below) of the synthetic circuit board  30  via the bonding wire W 5 . A more specific constitution of the matching circuit  50  will be described below. 
     Also in the matching circuits  50 , T-type filter circuits (matching circuits) are constituted of inductance components by the bonding wires W 4  and the bonding wires W 5 , and a capacitance of the metal pad  53  connected to between a node between these inductance components and the reference potential (bottom plate  4   e ). The matching circuits  50  achieve impedance matching with respect to the amplifier element portion  10  by performing impedance conversion through these T-type filter circuits. Normally, estimated impedances inside the transistors  13  from the drain pads  15  in the amplifier element portion  10  differ from the characteristic impedance (for example, 50Ω) of the transmission line and are generally values smaller than 50Ω. The matching circuits  50  cause the impedances to match an estimated impedance 50Ω inside the package  4  from the output terminal  3  through the T-type filter circuits. 
     The synthetic circuit board  30  is disposed on the bottom plate  4   e  of the package  4 . The synthetic circuit board  30  is disposed side by side with the amplifier element portion  10  and the output terminal  3  in the Y axis direction and is positioned between the amplifier element portion  10  and the output terminal  3 . The synthetic circuit board  30  has a ceramic substrate  31  and a synthetic circuit  32  provided on a main surface of the substrate  31 . For example, the planar shape of the substrate  31  is a rectangular shape. Along side  31   a  thereof on one side faces the amplifier element portion  10  via the matching circuits  50 , and a long side  31   b  thereof on the other side faces the output terminal  3 . A rear surface of the substrate  31  faces the bottom plate  4   e  of the package  4 . A short side  31   c  of the substrate  31  on one side is positioned in the vicinity of the side wall  4   c  of the package  4 , and a short side  31   d  of the substrate  31  on the other side is positioned in the vicinity of the side wall  4   d  of the package  4 . That is, the substrate  31  extends from an area in the vicinity of one end to an area in the vicinity of the other end of the package  4  in the X axis direction. 
     The synthetic circuit  32  synthesizes signals output from a plurality of drain pads  15  of the amplifier element portion  10  as one output signal. The synthetic circuit  32  includes a wiring pattern  33  provided on the main surface of the substrate  31 . The wiring pattern  33  has a line-symmetrical shape with respect to a centerline of the substrate  31  in the Y axis direction. The wiring pattern  33  includes four metal pads  33   a.  The four metal pads  33   a  are arrayed side by side along the long side  31   a.  Metal pads  33   a  adjacent to each other are connected to each other via a film resistor and constitute Wilkinson couplers. Accordingly, while isolation between the plurality of drain pads  15  of the amplifier element portion  10  is secured, matching of an output impedance of the amplifier element portion  10  viewed from the output terminal  3  is achieved. Each of the metal pads  33   a  is electrically connected to two corresponding metal pads  53  of the matching circuit  50  via the bonding wires W 5 . The wiring pattern  33  ultimately reaches a connection point with respect to the bonding wires W 6  while repeating coupling from the four metal pads  33   a.  The wiring pattern  33  is electrically connected to the output terminal  3  via the bonding wires W 6 . An amplified radio frequency signal is output to the output terminal  3  from a middle portion of the substrate  31  in the X axis direction. 
     Next, with reference to  FIG.  2   , the amplifier element  11  and the matching circuit  50  will be more specifically described.  FIG.  2    is a plan view illustrating the amplifier element  11  and the matching circuit  50  in  FIG.  1   . The amplifier element  11  includes a semiconductor substrate  12 , a plurality of transistors  13 , a plurality of gate pads  14 , a plurality of drain pads  15 , and a plurality of source pads  16 . The planar shape of the semiconductor substrate  12  is a rectangular shape having the X axis direction as a longitudinal direction. The plurality of transistors  13  are disposed side by side in the X axis direction on the semiconductor substrate  12 . Each of the numbers of gate pads  14 , drain pads  15 , and source pads  16  is the same as the number of transistors  13 . Each of the plurality of gate pads  14 , the plurality of drain pads  15 , and the plurality of source pads  16  is a metal film (for example, Au film) formed on a main surface of the semiconductor substrate  12 . 
     The plurality of gate pads  14  are electrically connected to the respective gate electrodes of the plurality of transistors  13 . The plurality of gate pads  14  are disposed side by side along an end side of each of the amplifier elements  11  on the input terminal  2  side. The plurality of drain pads  15  are electrically connected to the respective drain electrodes of the plurality of transistors  13 . The plurality of drain pads  15  are disposed side by side along an end side of each of the amplifier elements  11  on the output terminal  3  side. The plurality of source pads  16  are electrically connected to the respective source electrodes of the plurality of transistors  13 . The plurality of source pads  16  are alternately disposed side by side with the gate pads  14  along the end side of each of the amplifier elements  11  on the input terminal  2  side. Each of the source pads  16  is electrically connected to the bottom plate  4   e  of the package  4  via a via-hole penetrating the amplifier element  11  in a thickness direction (here, the Z axis direction) and serves as a reference potential. Each of the transistors  13  amplifies a radio frequency signal input to each of the gate pads  14  and outputs an amplified radio frequency signal from each of the drain pads  15 . 
     In  FIG.  2   , one amplifier element  11  is illustrated. In the present embodiment, one amplifier element  11  has a transistor group constituted of eight transistors  13 . Namely, in the high frequency amplifier  1 A, a plurality of transistor groups (two in the present embodiment) are provided on the bottom plate  4   e  of the package  4 . The eight transistors  13  include transistors  13 A,  13 B,  13 C,  13 D,  13 E,  13 F,  13 G, and  13 H which are arranged side by side in this order in the X axis direction. In other words, the transistor  13 D is arranged side by side with the transistor  13 A in the X axis direction, and the transistors  13 E,  13 F,  13 G, and  13 H are arranged side by side with the transistor  13 D on a side opposite to the transistor  13 A in the X axis direction. The transistor  13 A is an example of a first transistor according to the present embodiment, and the transistor  13 D is an example of a second transistor according to the present embodiment. Each of the transistors  13 E,  13 F,  13 G, and  13 H according to the present embodiment is an example of a third transistor. Each of the transistors  13 A and  13 H of the eight transistors  13  is disposed at an outermost part in the X axis direction. The transistor  13 B to the transistor  13 G are disposed between the transistors  13 A and  13 H. Namely, a different transistor  13  is adjacent to the transistors  13 A and  13 H on only one side thereof in the X axis direction. Different transistors  13  are adjacent to the transistor  13 B to the transistor  13 G on both sides thereof in the X axis direction. 
     Hereinafter, the drain pads  15  electrically connected to the respective drain electrodes of the transistor  13 A to the transistor  13 H will be respectively referred to as drain pads  15 A,  15 B,  15 C,  15 D,  15 E,  15 F,  15 G, and  15 H. The drain pad  15 A is an example of a first drain pad according to the present embodiment, and the drain pad  15 D is an example of a second drain pad according to the present embodiment. Each of the drain pads  15 E,  15 F,  15 G, and  15 H is an example of a third drain pad according to the present embodiment. 
     Hereinafter, the eight bonding wires W 4  corresponding to the drain pad  15 A to the drain pad  15 H will be respectively referred to as bonding wires W 41 , W 42 , W 43 , W 44 , W 45 , W 46 , W 47 , and W 48 . The bonding wire W 41  is an example of a first wire according to the present embodiment, and the bonding wire W 44  is an example of a second wire according to the present embodiment. Each of the bonding wires W 45 , W 46 , W 47 , and W 48  is an example of a third wire according to the present embodiment. The lengths of the eight bonding wires W 4  are equivalent to each other. Each of the bonding wires W 41  and W 48  of the eight bonding wires W 4  is disposed at an outermost part in the X axis direction. The bonding wire W 42  to the bonding wire W 47  are disposed between the bonding wires W 41  and W 48 . 
     In the present embodiment, one matching circuit  50  has two matching circuit patterns  52 . Each of the matching circuit patterns  52  has a plurality of transmission lines  54 . In the two matching circuit patterns  52 , the number of a plurality of transmission lines  54  is the same as the number of transistors  13 . In the present embodiment, one matching circuit pattern  52  has four transmission lines  54 . The four transmission lines  54  includes transmission lines  54 A,  54 B,  54 C, and  54 D. 
     The two matching circuit patterns  52  have mutually inverted constitutions with respect to the X axis direction. In the matching circuit pattern  52  on one side, the transmission lines  54 A,  54 B,  54 C, and  54 D corresponding to the transistors  13 A,  13 B,  13 C, and  13 D are arranged side by side in this order in the X axis direction. The transmission line  54 A in the matching circuit pattern  52  on one side is an example of a first transmission line according to the present embodiment. The transmission line  54 D in the matching circuit pattern  52  on one side is an example of a second transmission line according to the present embodiment. In the matching circuit pattern  52  on the other side, the transmission lines  54 D,  54 C,  54 B, and  54 A corresponding to the transistors  13 E,  13 F,  13 G, and  13 H are arranged side by side in this order in the X axis direction. Each of the transmission lines  54 D,  54 C,  54 B, and  54 A in the matching circuit pattern  52  on the other side is an example of a third transmission line according to the present embodiment. In the eight transmission lines  54 , each of two transmission lines  54 A is disposed at an outermost part in the X axis direction. Each of the transmission lines  54 B,  54 C, and  54 D is disposed between the two transmission lines  54 A. 
     The transmission lines  54 A and  54 B are constituted of a metal pad  53 A (first pad) which is the metal pad  53  described above. The transmission lines  54 C and  54 D are constituted of a metal pad  53 B (second pad) which is the metal pad  53  described above. The metal pad  53 A is an example of the first pad according to the present embodiment. The metal pad  53 B is an example of the second pad according to the present embodiment. The metal pad  53 A and the metal pad  53 B are connected to each other via a film resistor  55   a.  The width (here, the largest dimension in the X axis direction) of the metal pad  53 A and the width (here, the largest dimension in the X axis direction) of the metal pad  53 B are equivalent to each other. The length (here, the largest dimension in the Y axis direction) of the metal pad  53 A and the length (here, the largest dimension in the Y axis direction) of the metal pad  53 B are equivalent to each other. 
       FIG.  3    is an enlarged plan view of one matching circuit pattern  52  in  FIG.  2   . An end portion of the metal pad  53 A on one side in the Y axis direction branches into an input end  53   a  in the transmission line  54 A and an input end  53   b  in the transmission line  54 B. The transmission line  54 A is connected to the bonding wire W 41  or the bonding wire W 48  at a connection point P 1  (first connection point) in the input end  53   a.  The transmission line  54 B is connected to the bonding wire W 42  or the bonding wire W 47  at a connection point P 2  in the input end  53   b.  The input end  53   a  and the input end  53   b  are connected to each other via a film resistor  55   b.  The transmission line  54 A and the transmission line  54 B are coupled to a connection point Q 1  at an end portion of the metal pad  53 A on the other side in the Y axis direction and are connected to the bonding wire W 5 . 
     An end portion of the metal pad  53 B on one side in the Y axis direction branches into an input end  53   c  in the transmission line  54 C and an input end  53   d  in the transmission line  54 D. The transmission line  54 C is connected to the bonding wire W 43  or the bonding wire W 46  at a connection point P 3  in the input end  53   c.  The transmission line  54 D is connected to the bonding wire W 44  or the bonding wire W 45  at a connection point P 4  (second connection point) in the input end  53   d.  The input end  53   c  and the input end  53   d  are connected to each other via a film resistor  55   c.  The transmission line  54 C and the transmission line  54 D are coupled to a connection point Q 2  at an end portion of the metal pad  53 B on the other side in the Y axis direction and are connected to the bonding wire W 5 . 
     The external shape of the metal pad  53 A and the external shape of the metal pad  53 B are asymmetrical with each other with respect to a virtual straight line N 1  (second virtual straight line) which is orthogonal to a virtual straight line (first virtual straight line) connecting the connection point P 1  and the connection point P 4  to each other and passes through a median point between the connection point P 1  and the connection point P 4 . In other words, the matching circuit pattern  52  has an asymmetrical external shape with respect to the virtual straight line N 1 . Due to such an external shape, the length of a part of the virtual straight line N 1  on one side (here, a side including the connection point P 1 ) at an outer edge of the matching circuit pattern  52  is longer than the length of a part of the virtual straight line N 1  on the other side (here, a side including the connection point P 4 ). Specifically, the length of a part of the virtual straight line N 1  on one side is a length of a part having a position projected in the X axis direction from the connection point P 1  as a starting point and a position projected in the Y axis direction from the connection point Q 1  as an ending point at the outer edge of the matching circuit pattern  52  including the metal pad  53 A. The length of a part of the virtual straight line N 1  on the other side is a length of a part having a position projected in the X axis direction from the connection point P 4  as a starting point and a position projected in the Y axis direction from the connection point Q 2  as an ending point at the outer edge of the matching circuit pattern  52  including the metal pad  53 B. 
     The metal pad  53 A exhibits substantially a rectangular shape in a plan view and has four corner portions. The four corner portions include one chamfered corner portion C 1  (first corner portion). In the four corner portions, three corner portions excluding the corner portion C 1  are not chamfered. The metal pad  53 A is connected to the metal pad  53 B on a long side  53   r  on one side. The corner portion C 1  is positioned at an intersection portion between a long side  53   s  on the other side in the metal pad  53 A and a short side  53   t  facing the synthetic circuit board  30  (refer to  FIG.  1   ). The transmission line  54 A is constituted of a part including the corner portion C 1  in the metal pad  53 A. The transmission line  54 B is constituted of a part not including the corner portion C 1  in the metal pad  53 A. 
     The metal pad  53 B exhibits substantially a rectangular shape in a plan view and has four corner portions. The four corner portions include one chamfered corner portion C 2  (second corner portion). The corner portion C 2  is chamfered by a chamfering amount L 2  larger than a chamfering amount L 1  of the corner portion C 1 . For example, the chamfering amount L 2  is approximately three times the chamfering amount L 1 . In the four corner portions, three corner portions excluding the corner portion C 2  are not chamfered. The metal pad  53 B is connected to the metal pad  53 A on a long side  53   u  on one side. The corner portion C 2  is positioned at an intersection portion between a long side  53   v  on the other side in the metal pad  53 B and a short side  53   w  facing the synthetic circuit board  30  (refer to  FIG.  1   ). The transmission line  54 C is constituted of a part not including the corner portion C 2  in the metal pad  53 B. The transmission line  54 D is constituted of a part including the corner portion C 2  in the metal pad  53 B. Therefore, the length of a part constituting the transmission line  54 D at an outer edge of the metal pad  53 B is shorter than the length of a part constituting the transmission line  54 A at an outer edge of the metal pad  53 A. 
     Next, an electrical length in the constitution from the amplifier element  11  to the synthetic circuit  32  will be described.  FIG.  4    is an explanatory perspective view of each electrical length from the bonding wire W 41  to the bonding wire W 48 . As illustrated in  FIG.  4   , the bonding wire W 41  mainly receives an influence of magnetic field coupling from the closest bonding wire W 42  and an influence of magnetic field coupling from the second closest bonding wire W 43 . In contrast, the bonding wire W 44  mainly receives influences of magnetic field coupling from each of the closest bonding wires W 43  and W 45  and magnetic field coupling from the second closest bonding wires W 42  and W 46 . Accordingly, mutual inductance components of the bonding wire W 41  increase in accordance with an influence from only one side, whereas mutual inductance components of the bonding wire W 44  increase in accordance with influences from both sides. Namely, mutual inductance components of each of the bonding wire W 41  to the bonding wire W 48  increase in accordance with a different bonding wire W 4  close thereto. 
       FIG.  5    is a Smith chart showing each impedance of the bonding wire W 41  to the bonding wire W 48 .  FIG.  5    illustrates an S parameter (S 11 ) for the transmission line terminal corresponding to the impedance of each of the bonding wire W 41  to the bonding wire W 48  within a frequency range of 10.700 GHz to 12.700 GHz. The impedance of each of the bonding wire W 41  to the bonding wire W 48  means an effective impedance of each of the bonding wire W 41  to the bonding wire W 48 . The effective impedance means an impedance in one bonding wire W 4  in consideration of an influence from a different bonding wire W 4 . In the Smith chart of  FIG.  5   , the range is displayed up to a reflection coefficient Γ=0.3, and the impedance 50Ω at the center of the Smith chart is standardized at the characteristic impedance Z 0 =50Ω of the calculation port. Since the value of the inductance increases in the direction of the arrow in  FIG.  5   , it is ascertained from  FIG.  5    that the inductances of the bonding wires W 44  and W 45  are larger than the bonding wires W 41  and W 48 . 
       FIG.  6    is a graph showing each inductance of the bonding wire W 41  to the bonding wire W 48 . In the graph of  FIG.  6   , the horizontal axis indicates a frequency, and the vertical axis indicates a value of the inductance. It is also ascertained from  FIG.  6    that the inductances of the bonding wires W 44  and W 45  are larger than the inductances of the bonding wires W 41  and W 48 . In this manner, in the plurality of bonding wires W 4  arranged side by side in a row, the inductance increases toward the center of the row, and the inductance decreases toward the end of the row. Therefore, the effective electrical lengths of the bonding wire W 41  to the bonding wire W 48  become longer in the wire closer to the center of the row and become shorter in the wire closer to the end of the row. Specifically, the effective electrical length of the bonding wire W 44  is longer than the effective electrical length of the bonding wire W 41 . The effective electrical length means an electrical length of a predetermined path through which electromagnetic waves pass and an electrical length in consideration of an influence of mutual inductances from a bonding wire W 4  different of the bonding wire W 4  included in the path. 
     Next, the electrical lengths of the transmission lines  54 A and  54 D will be described. Since a radio frequency signal is propagated through the transmission lines  54 A and  54 D as an electromagnetic wave, the electrical lengths of the transmission lines  54 A and  54 D correspond to the length of the path through which electromagnetic waves pass in the transmission lines  54 A and  54 D. 
     First, with reference to  FIGS.  7  and  8   , a relationship between a length of a physical path through which electromagnetic waves pass and a phase will be described.  FIGS.  7  and  8    are explanatory views of a relationship between a length of a physical path through which electromagnetic waves pass and a phase.  FIG.  7    illustrates a path D 1  through which an electromagnetic wave E passes, and  FIG.  8    illustrates a path D 2  through which the electromagnetic wave E passes. The path D 2  is longer than the path D 1 . In the examples of  FIGS.  7  and  8   , the length of the path D 2  is approximately 1.4 times the length of the path D 1 . Accordingly, a phase difference occurs between the electromagnetic wave E which has passed through the path D 1  and the electromagnetic wave E which has passed through the path D 2 . Here, the phase of the electromagnetic wave E at a terminal of the path D 1  is 0 deg, whereas the phase of the electromagnetic wave E at a terminal of the path D 2  is approximately 90 deg. Therefore, a phase difference of approximately 90 deg has occurred. 
     Here, a path through which a radio frequency (microwave) which is an electromagnetic wave passes will be specifically described.  FIG.  9    is an explanatory view of a distribution state of currents in microwaves. (a) portion of  FIG.  9    is a cross-sectional view illustrating a dielectric substrate  90  and a metal conductor  80  provided on the dielectric substrate  90 , and (b) portion of  FIG.  9    is a graph showing a magnitude of a current flowing in the Y axis direction through the metal conductor  80 . For example, the metal conductor  80  is a microstrip conductor. In the graph shown in  FIG.  9   , the horizontal axis indicates a position in the X axis direction having the center of the metal conductor  80  in the X axis direction as zero, and the vertical axis indicates a current value I. 
       FIG.  10    is a view of a simulation illustrating a concentration state of currents in microwaves.  FIG.  10    illustrates a state in which more currents are concentrated at a darker part. From  FIG.  10   , it is ascertained that currents are particularly in a concentrated state at both end portions of the metal conductor  80  in the X axis direction. Therefore, as illustrated in  FIG.  9   , regarding microwaves, when a current flows in the metal conductor  80 , it is ascertained that the current value I increases at both end portions of the metal conductor  80  in the X axis direction. 
       FIG.  11    is a view of a simulation illustrating a distribution state of electric field intensities in microwaves.  FIG.  11    illustrates that the electric field intensity increases at a darker part. From  FIG.  11   , it is ascertained that the electric field intensity particularly increases at both end portions of the metal conductor  80  in the X axis direction. Therefore, it is ascertained that microwaves which are electromagnetic waves mainly pass through an outer edge of a pattern such as the foregoing metal conductor  80 . In other words, a physical path through which electromagnetic waves pass corresponds to an outer edge of a pattern. Therefore, the length of the physical path through which electromagnetic waves pass changes by changing the length of the outer edge of the pattern. The length of the outer edge of the pattern can be changed by changing the shape of the outer edge. 
       FIG.  12    is a plan view illustrating one reference example of a pattern  80 A through which electromagnetic waves pass, and  FIG.  13    is a plan view illustrating another reference example of a pattern  80 B through which electromagnetic waves pass. Both the patterns  80 A and  80 B have two parallel input ports  81  and  82  and one output port  83 . Each of the patterns  80 A and  80 B exhibits substantially a rectangular shape and is bifurcated into a part constituting the input port  81  and a part constituting the input port  82  at one end portion in the longitudinal direction. The output port  83  is positioned at the other end portion in the longitudinal direction of the patterns  80 A and  80 B. The distance from the input port  81  to the output port  83  and the distance from the input port  82  to the output port  83  are equivalent to each other. 
     Here, the external shape of the pattern  80 A is line-symmetrical with respect to a virtual straight line N 82  which is orthogonal to a virtual straight line N 81  connecting the input ports  81  and  82  to each other and passes through a median point between the input ports  81  and  82 . In contrast, the external shape of the pattern  80 B is asymmetrical with respect to the virtual straight line N 82 . In the examples of  FIGS.  12  and  13   , all four corner portions in the pattern  80 A form right angles, whereas four corner portions in the pattern  80 B include three corner portions forming right angles and one chamfered corner portion. Specifically, the external shape of the pattern  80 B exhibits a shape in which a corner portion forming a right angle is cut in a portion of a part from the input port  81  to the output port  83 . For this reason, at an outer edge of the pattern  80 B, the length from the input port  81  to the output port  83  is shorter than the length from the input port  82  to the output port  83 . 
     As described above, since microwaves which are electromagnetic waves mainly pass through the outer edges of the patterns  80 A and  80 B, a path DA from the input port  82  to the output port  83  is shortcut with respect to a path DB from the input port  81  to the output port  83 . In this manner, when a physical length difference occurs between the paths DA and DB, as described above, a phase difference occurs between an electromagnetic wave which has passed through the path DA and an electromagnetic wave which has passed through the path DB. 
       FIG.  14    is a graph showing a phase difference occurring between two electromagnetic waves which have respectively passed through the paths DA and DB in the patterns  80 A and  80 B. In the graph of  FIG.  14   , the horizontal axis indicates the frequency, and the vertical axis indicates the phase difference. From  FIG.  14   , it is ascertained that no phase difference has occurred in the pattern  80 A but a phase difference has occurred in the pattern  80 B. In  FIG.  14   , since the absolute value of the phase difference occurring in the pattern  80 B becomes large as the frequency increases, it is ascertained that an influence on the phase difference occurring due to change in physical length of the paths DA and DB becomes significant as the frequency increases. 
     With reference to  FIG.  3    again based on those above, the electrical lengths of the transmission lines  54 A and  54 D according to the present embodiment will be described. As described above, the length of a part constituting the transmission line  54 D at the outer edge of the metal pad  53 B is shorter than the length of a part constituting the transmission line  54 A at the outer edge of the metal pad  53 A. Therefore, the electrical length of the transmission line  54 D is shorter than the electrical length of the transmission line  54 A. 
     In the present embodiment, for example, the absolute value of the electrical length difference between the transmission lines  54 A and  54 D is substantially equivalent to the absolute value of the electrical length difference between the bonding wires W 41  and W 44  described above. In this specification, the expression “substantially equivalent” means that a difference between two values is small enough such that it can be disregarded. Accordingly, the total electrical length of the effective electrical length of the bonding wire W 41  and the electrical length of the transmission line  54 A has become substantially equivalent to the total electrical length of the effective electrical length of the bonding wire W 44  and the electrical length of the transmission line  54 D. In other words, the effective electrical length from the drain pad  15 A to the wiring pattern  33  is substantially equivalent to the effective electrical length from the drain pad  15 D to the wiring pattern  33 . In the present embodiment, the effective electrical length difference occurring between the bonding wires W 41  and W 44  and the electrical length difference occurring between the transmission lines  54 A and  54 D are mutually offset. Accordingly, no phase difference occurs between an electromagnetic wave which has passed through the path from the drain pad  15 A to the wiring pattern  33  and an electromagnetic wave which has passed through the path from the drain pad  15 D to the wiring pattern  33 . 
     Advantageous effects of the foregoing high frequency amplifier  1 A will be described. In this high frequency amplifier  1 A, each of the plurality of transistors  13  is connected to the matching circuit pattern  52  through the bonding wire W 4 . In this constitution, variation may occur in the effective impedance between the plurality of bonding wires W 4  in accordance with the degree of an influence received from a different bonding wire W 4 . Such variation in effective impedance between the plurality of bonding wires W 4  is manifested as variation in effective electrical length between the plurality of bonding wires W 4 . Specifically, the effective electrical length of the bonding wire W 44  is longer than the effective electrical length of the bonding wire W 41 . Here, in this high frequency amplifier  1 A, since the matching circuit pattern  52  has an asymmetrical external shape, the electrical length of the transmission line  54 D is shorter than the electrical length of the transmission line  54 A. Accordingly, in a constitution in which the effective electrical length of the bonding wire W 44  is longer than the effective electrical length of the bonding wire W 41 , the electrical length difference between the bonding wires W 41  and W 44  may be canceled out by the electrical length difference between the transmission lines  54 A and  54 D. Therefore, variation between the effective electrical length from the transistor  13 A (more specifically, the drain pad  15 A) to the wiring pattern  33  and the effective electrical length from the transistor  13 D (more specifically, the drain pad  15 D) to the wiring pattern  33  is reduced. Since variation in effective impedance between the plurality of bonding wires W 4  is reduced when variation in electrical length between the plurality of bonding wires W 4  is reduced, variation in phase between the transistors  13 A and  13 D can be reduced due to this constitution. The same also applies to that between the transistors  13 E and  13 H. 
     In the foregoing high frequency amplifier  1 A, the matching circuit pattern  52  has the metal pad  53 A including the corner portion C 1  and the metal pad  53 B including the corner portion C 2 . The transmission line  54 A is constituted of a part including the corner portion C 1  in the metal pad  53 A. The transmission line  54 D is constituted of a part including the corner portion C 2  in the metal pad  53 B. The corner portion C 2  is chamfered by the chamfering amount L 2  larger than the chamfering amount L 1  of the corner portion C 1 . According to knowledge of the inventors, as described above, currents (electric field) in a radio frequency is likely to be concentrated at an outer edge of a conductor. For this reason, in the transmission line  54 A, a radio frequency signal is propagated along the corner portion C 1  in the metal pad  53 A. In the transmission line  54 D, a radio frequency signal is propagated along the corner portion C 2  in the metal pad  53 B. Here, since the corner portion C 2  is chamfered by the chamfering amount L 2  larger than the chamfering amount L 1  of the corner portion C 1 , the path including the corner portion C 2  becomes shorter than the path including the corner portion C 1 , and thus a constitution in which the electrical length of the transmission line  54 D is shorter than the electrical length of the transmission line  54 A may be realized. According to this constitution, the effective electrical length from the transistor  13 A to the wiring pattern  33  and the effective electrical length from the transistor  13 D to the wiring pattern  33  are likely to be subjected to matching through adjustment of the chamfering amounts L 1  and L 2 . Therefore, variation in phase between the transistors  13 A and  13 D can be more reliably reduced. The same also applies to that between the transistors  13 E and  13 H. 
     In the foregoing high frequency amplifier  1 A, a plurality of transistor groups configured to include the transistor  13 A, the transistor  13 D, and the transistors  13 E,  13 F,  13 G, and  13 H are provided. Due to this constitution, the high frequency amplifier  1 A having a high output can be realized. 
     In the foregoing high frequency amplifier  1 A, the bonding wire W 41  is disposed at the outermost part of the transistor groups, and the bonding wire W 44  is disposed between the bonding wire W 41  and the bonding wires W 45 , W 46 , W 47 , and W 48 . The length of the bonding wire W 41  is equivalent to the length of the bonding wire W 44 , and the effective electrical length from the drain pad  15 A to the wiring pattern  33  is substantially equivalent to the effective electrical length from the drain pad  15 D to the wiring pattern  33 . Therefore, the phases substantially coincide with each other between the transistors  13 A and  13 D. The same also applies to that between the transistors  13 E and  13 H. Hence, this is particularly advantageous from the viewpoint of improving the electric power efficiency of the high frequency amplifier  1 A. 
     Next, the high frequency amplifier  1 B according to the second embodiment will be described. As described above,  FIG.  1    is a plan view illustrating an internal constitution of the high frequency amplifier  1 A according to the first embodiment and the high frequency amplifier  1 B according to the second embodiment. The high frequency amplifier  1 B differs from the high frequency amplifier  1 A in including a matching circuit  60  in place of the matching circuit  50  and is otherwise the same as the high frequency amplifier  1 A in constitution. In the present embodiment, the high frequency amplifier  1 B includes two matching circuits  60  as an example. Hereinafter, points differing from the high frequency amplifier  1 A will be described. 
       FIG.  15    is a plan view illustrating the matching circuit  60 . The matching circuit  60  differs from the matching circuit  50  in including two matching circuit patterns  62  in place of two matching circuit patterns  52  and is otherwise the same as the matching circuit  50  in constitution. Each of the matching circuit patterns  62  has a plurality of metal pads  63 . Similar to the metal pads  53 , each of the metal pads  63  is electrically connected to the corresponding drain pad  15  of the amplifier element portion  10  via the bonding wire W 4  and is electrically connected to the corresponding metal pad  33   a  of the synthetic circuit board  30  via the bonding wire W 5 . 
     The matching circuit pattern  62  has a plurality of transmission lines  64 . In the two matching circuit patterns  62 , the number of a plurality of transmission lines  64  is the same as the number of transistors  13  (refer to  FIG.  2   ). One matching circuit pattern  62  has four transmission lines  64 . The four transmission lines  64  include transmission lines  64 A,  64 B,  64 C, and  64 D. 
     The two matching circuit patterns  62  have mutually inverted constitutions with respect to the X axis direction. In the matching circuit pattern  62  on one side, the transmission lines  64 A,  64 B,  64 C, and  64 D corresponding to the transistors  13 A,  13 B,  13 C, and  13 D (refer to  FIG.  2   ) are arranged side by side in this order in the X axis direction. The transmission line  64 A in the matching circuit pattern  62  on one side is an example of the first transmission line according to the present embodiment. The transmission line  64 D in the matching circuit pattern  62  on one side is an example of the second transmission line according to the present embodiment. The transmission lines  64 B and  64 C in the matching circuit pattern  62  on one side may be examples of the second transmission line according to the present embodiment. In the matching circuit pattern  62  on the other side, the transmission lines  64 D,  64 C,  64 B, and  64 A corresponding to the transistors  13 E,  13 F,  13 G, and  13 H are arranged side by side in this order in the X axis direction. The transmission lines  64 D,  64 C,  64 B, and  64 A in the matching circuit pattern  62  on the other side are examples of the third transmission line according to the present embodiment. In eight transmission lines  64 , each of the two transmission lines  64 A is disposed at the outermost part in the X axis direction. Each of the transmission lines  64 B,  64 C, and  64 D is disposed between the two transmission lines  64 A. 
     The transmission lines  64 A and  64 B are constituted of a metal pad  63 A which is the metal pad  63  described above. The transmission lines  64 C and  64 D are constituted of a metal pad  63 B which is the metal pad  63 . The metal pad  63 A is an example of the first pad according to the present embodiment. The metal pad  63 B is an example of the second pad according to the present embodiment. The metal pad  63 A and the metal pad  63 B are connected to each other via a film resistor  65   a.  The width (here, the largest dimension in the X axis direction) of the metal pad  63 A is smaller than the width (here, the largest dimension in the X axis direction) of the metal pad  63 B. The length (here, the largest dimension in the Y axis direction) of the metal pad  63 A and the length (here, the largest dimension in the Y axis direction) of the metal pad  63 B are equivalent to each other. 
       FIG.  16    is an enlarged plan view of one matching circuit pattern  62  in  FIG.  15   . An end portion of the metal pad  63 A on one side in the Y axis direction branches into an input end  63   a  in the transmission line  64 A and an input end  63   b  in the transmission line  64 B. The transmission line  64 A is connected to the bonding wire W 41  or the bonding wire W 48  at a connection point P 5  (first connection point) in the input end  63   a.  The transmission line  64 B is connected to the bonding wire W 42  or the bonding wire W 47  at a connection point P 6  in the input end  63   b.  The input end  63   a  and the input end  63   b  are connected to each other via a film resistor  65   b.  The transmission line  64 A and the transmission line  64 B are coupled to a connection point Q 3  at an end portion of the metal pad  63 A on the other side in the Y axis direction and are connected to the bonding wire W 5 . 
     An end portion of the metal pad  63 B on one side in the Y axis direction branches into an input end  63   c  in the transmission line  64 C and an input end  63   d  in the transmission line  64 D. The transmission line  64 C is connected to the bonding wire W 43  or the bonding wire W 46  at a connection point P 7  in the input end  63   c.  The transmission line  64 D is connected to the bonding wire W 44  or the bonding wire W 45  at a connection point P 8  (second connection point) in the input end  63   d.  The input end  63   c  and the input end  63   d  are connected to each other via a film resistor  65   c.  The transmission line  64 C and the transmission line  64 D are coupled to a connection point Q 4  at an end portion of the metal pad  63 B on the other side in the Y axis direction and are connected to the bonding wire W 5 . 
     The external shape of the metal pad  63 A and the external shape of the metal pad  63 B are asymmetrical with each other with respect to a virtual straight line N 2  (second virtual straight line) which is orthogonal to a virtual straight line (first virtual straight line) connecting the connection point P 5  and the connection point P 8  to each other and passes through a median point between the connection point P 5  and the connection point P 8 . In other words, the matching circuit pattern  62  has an asymmetrical external shape with respect to the virtual straight line N 2 . Due to such an external shape, at an outer edge of the matching circuit pattern  62 , the length of a part of the virtual straight line N 2  on one side (here, a side including the connection point P 5 ) is longer than the length of a part of the virtual straight line N 2  on the other side (here, a side including the connection point P 8 ). Specifically, the length of a part of the virtual straight line N 2  on one side is a length of a part having a position projected in the X axis direction from the connection point P 5  as a starting point and a position projected in the Y axis direction from the connection point Q 3  as an ending point at the outer edge of the matching circuit pattern  62  including the metal pad  63 A. The length of a part of the virtual straight line N 2  on the other side is a length of a part having a position projected in the X axis direction from the connection point P 8  as a starting point and a position projected in the Y axis direction from the connection point Q 4  as an ending point at the outer edge of the matching circuit pattern  62  including the metal pad  63 B. 
     Similar to the metal pad  53 A, the metal pad  63 A exhibits substantially a rectangular shape in a plan view and has four corner portions including one corner portion C 1  (first corner portion). The metal pad  63 A is connected to the metal pad  63 B on a long side  63   r  on one side. The corner portion C 1  is positioned at an intersection portion between a long side  63   s  on the other side in the metal pad  63 A and a short side  63   t  facing the synthetic circuit board  30  (refer to  FIG.  1   ). The transmission line  64 A is constituted of a part including the corner portion C 1  in the metal pad  63 A. The transmission line  64 B is constituted of a part not including the corner portion C 1  in the metal pad  63 A. 
     Similar to the metal pad  53 B, the metal pad  63 B exhibits substantially a rectangular shape in a plan view and has four corner portions including one corner portion C 2  (second corner portion). The metal pad  63 B is connected to the metal pad  63 A on one side on a long side  63   u.  The corner portion C 2  is positioned at an intersection portion between a long side  63   v  on the other side in the metal pad  63 B and a short side  63   w  facing the synthetic circuit board  30  (refer to  FIG.  1   ). The transmission line  64 C is constituted of a part not including the corner portion C 2  in the metal pad  63 B. The transmission line  64 D is constituted of a part including the corner portion C 2  in the metal pad  63 B. 
     In the present embodiment, the length of a part constituting the transmission line  64 D at an outer edge of the metal pad  63 B is shorter than the length of a part constituting the transmission line  64 A at an outer edge of the metal pad  63 A. As an example, in the metal pad  63 A, a length L 3  of the long side  63   s  is approximately 485 μm, a length L 4  of the short side  63   t  is approximately 325 μm, and a length L 5  of a chamfered portion in the corner portion C 1  is approximately 78 μm. The length of a part constituting the transmission line  64 A at the outer edge of the metal pad  63 A (that is, the total value of the lengths L 3 , L 4 , and L 5 ) is approximately 888 μm herein. In the metal pad  63 B, a length L 6  of the long side  63   v  is approximately 360 μm, a length L 7  of the short side  63   w  is approximately 255 μm, and a length L 8  of a chamfered portion in the corner portion C 2  is 255 μm. The length of a part constituting the transmission line  64 D at the outer edge of the metal pad  63 B (that is, the total value of the lengths L 6 , L 7 , and L 8 ) is approximately 870 μm herein. 
     The metal pad  63 A has an asymmetrical external shape with respect to a virtual straight line N 3  (second virtual straight line) which is orthogonal to a virtual straight line (first virtual straight line) connecting the connection point P 5  and the connection point P 6  to each other and passes through a median point between the connection point P 5  and the connection point P 6 . The metal pad  63 B has an asymmetrical external shape with respect to a virtual straight line N 4  (second virtual straight line) which is orthogonal to a virtual straight line (first virtual straight line) connecting the connection point P 7  and the connection point P 8  to each other and passes through a median point between the connection point P 7  and the connection point P 8 . Specifically, a width L 11  (here, the dimension in the X axis direction) of the input end  63   a  and a width L 12  (here, the dimension in the X axis direction) of the input end  63   b  are different from each other, and a width L 13  (here, the dimension in the X axis direction) of the input end  63   c  and a width L 14  (here, the dimension in the X axis direction) of the input end  63   d  are different from each other. The width L 11  is smaller than the width L 12 . That is, the transmission line  64 A is thinner than the transmission line  64 B. The width L 13  is smaller than the width L 14 . That is, the transmission line  64 C is thinner than the transmission line  64 D. 
     In the present embodiment, the transmission lines  64 A and  64 B are thinner than the transmission lines  64 C and  64 D. The widths L 11 , L 12 , L 13 , and L 14  are different from each other and are larger in this order. In other words, in a plurality of input ends arranged side by side in the X axis direction, an input end closer to the center of the dielectric substrate  51  in the X axis direction has a larger width (here, the dimension in the X axis direction). For example, the width L 11  is approximately 0.6 times the width L 14 , the width L 12  is approximately 0.85 times the width L 14 , and the width L 13  is approximately 0.9 times the width L 14 . As an example, the width L 11  is approximately 120 μm, the width L 12  is approximately 170 μm, the width L 13  is approximately 180 μm, and the width L 14  is approximately 200 μm. In this example, in consideration of the thickness (specifically, 200 μm) of the dielectric substrate  51  described above and the relative dielectric constant (specifically, εr=150), when the frequency is 11.7 GHz, the effective wavelength in the transmission line  64 A becomes 123.4 deg, and the effective wavelength in the transmission line  64 D becomes 120.9 deg. 
     The electrical lengths of the transmission lines  64 A and  64 D will be described. As described above, the length of a part constituting the transmission line  64 D at the outer edge of the metal pad  63 B is shorter than the length of a part constituting the transmission line  64 A at the outer edge of the metal pad  63 A. Moreover, in the present embodiment, since the transmission line  64 A is thinner than the transmission line  64 D, the impedance of the transmission line  64 A is larger than the impedance of the transmission line  64 D. Therefore, the electrical length of the transmission line  64 D is shorter than the electrical length of the transmission line  64 A. 
     In the present embodiment, for example, the absolute value of the electrical length difference between the transmission lines  64 A and  64 D is equivalent to the absolute value of the effective electrical length difference between the bonding wires W 41  and W 44  described above. Accordingly, the total electrical length of the effective electrical length of the bonding wire W 41  and the electrical length of the transmission line  64 A has become substantially equivalent to the total electrical length of the effective electrical length of the bonding wire W 44  and the electrical length of the transmission line  64 D. In other words, the effective electrical length from the drain pad  15 A to the wiring pattern  33  is substantially equivalent to the effective electrical length from the drain pad  15 D to the wiring pattern  33 . In the present embodiment, the effective electrical length difference occurring between the bonding wires W 41  and W 44  and the electrical length difference occurring between the transmission lines  64 A and  64 D are mutually offset. Accordingly, no phase difference occurs between an electromagnetic wave which has passed through the path from the drain pad  15 A to the wiring pattern  33  and an electromagnetic wave which has passed through the path from the drain pad  15 D to the wiring pattern  33 . 
     Advantageous effects of the foregoing high frequency amplifier  1 B will be described. First, a comparative example will be described. A high frequency amplifier according to the comparative example differs from the high frequency amplifier  1 B in including a matching circuit  60 X according to the comparative example in place of the matching circuit  60 .  FIG.  19    is a plan view illustrating the matching circuit  60 X of the high frequency amplifier according to the comparative example. The matching circuit  60 X differs from the matching circuit  60  in including two matching circuit patterns  62 X in place of the two matching circuit patterns  62  and is otherwise the same as the matching circuit  60  in constitution. 
     Each of the matching circuit patterns  62 X has two metal pads  63 X in place of the metal pads  63 A and  63 B. The metal pads  63 X differ from the metal pads  63 A and  63 B in having no chamfered corner portions C 1  and C 2 . In the metal pads  63 X, all four corner portions are constituted to form right angles. Two metal pads  63 X in the matching circuit patterns  62 X differ from the metal pads  63 A and  63 B in having four input ends  63   y  in place of the input ends  63   a,    63   b ,  63   c,  and  63   d.  The widths (here, the dimensions in the X axis direction) of the four input ends  63   y  are equivalent to each other. The metal pads  63 X are otherwise the same as the metal pads  63 A and  63 B in constitution. Namely, the external shapes of the matching circuit patterns  62 X according to the comparative example are line-symmetrical with respect to the virtual straight line N 2 . 
       FIG.  20    is a Smith chart showing each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D of the high frequency amplifier according to the comparative example. In  FIG.  20   , each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D is indicated within a frequency range of 10.700 GHz to 12.700 GHz. From  FIG.  20   , it is ascertained that a range of variation in phase occurring between the transistors  13 A,  13 B,  13 C, and  13 D (here, a phase difference ΔZ 1  occurring between the transistors  13 A and  13 D) becomes large. In the example of  FIG.  20   , the phase difference ΔZ 1  is approximately 9 deg. 
     It is considered that each load impedance in the transistors  13 H,  13 G,  13 F, and  13 E is approximately the same as each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D. Hence, in the high frequency amplifier according to the comparative example, it is ascertained that significant variation has occurred in phase between the transistor  13 A to the transistor  13 H (in the example of  FIG.  20   , variation within a range of approximately 9 deg). 
       FIG.  21    is a view illustrating a relationship between variation in phase of a load impedance and electric power efficiency according to the comparative example. The contour diagram of  FIG.  21    illustrates that the electric power efficiency becomes higher at a darker part. The Smith chart of  FIG.  21    illustrates a peak point Z 10  of the phase of a load impedance in which the maximum electric power efficiency (here, ηd=52.9%) can be obtained at a frequency of 11.700 GHz. The peak point Z 10  is 99 deg herein. From  FIG.  21   , at a point Z 11  (here, 108 deg) of which the phase is shifted by approximately 9 deg with respect to the peak point Z 10 , it is ascertained that the electric power efficiency becomes lower (here, approximately ηd=50.0%) than that at the peak point Z 10 . Therefore, when the matching circuit  60 X is used, it is ascertained that the electric power efficiency becomes lower than that at the peak point Z 10  by approximately 3%. 
     In contrast, in the high frequency amplifier  1 B according to the present embodiment, due to a constitution similar to that of the high frequency amplifier  1 A described above, effects similar to those of the high frequency amplifier  1 A can be achieved. Moreover, in the high frequency amplifier  1 B, the transmission line  64 A is thinner than the transmission line  64 D. Accordingly, since the impedance per unit length of the transmission line  64 A can become higher than the impedance per unit length of the transmission line  64 D, a constitution in which the electrical length of the transmission line  64 D is shorter than the electrical length of the transmission line  64 A may be realized. According to the degree of thinning of the transmission line  64 A, the effective electrical length from the transistor  13 A to the wiring pattern  33  and the effective electrical length from the transistor  13 D to the wiring pattern  33  are likely to be subjected to matching. Therefore, variation in phase between the transistor  13 A and the transistor  13 D can be more reliably reduced. Due to a similar reason, variation in phase between the transistor  13 A to the transistor  13 H can be more reliably reduced. 
       FIG.  17    is a Smith chart showing each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D of the high frequency amplifier  1 B according to the second embodiment. In  FIG.  17   , each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D is indicated within a frequency range of 10.700 GHz to 12.700 GHz. From  FIG.  17   , it is ascertained that a range of variation in phase occurring between the transistors  13 A,  13 B,  13 C, and  13 D (here, a phase difference ΔZ 2  occurring between the transistors  13 A and  13 D) is small. In the example of  FIG.  17   , the phase difference ΔZ 2  is approximately 4 deg. 
     It is considered that each load impedance in the transistors  13 H,  13 G,  13 F, and  13 E is approximately the same as each load impedance in the transistors  13 A,  13 B,  13 C, and  13 D. Hence, according to the high frequency amplifier  1 B, it is ascertained that only small variation has occurred in phase between the transistor  13 A to the transistor  13 H (in the example of  FIG.  17   , variation within a range of approximately 4 deg). For this reason, it is ascertained that variation in phase occurring between the transistor  13 A to the transistor  13 H can be further reduced in the high frequency amplifier  1 B than in the high frequency amplifier according to the comparative example. In the example of  FIG.  17   , it is ascertained that the range of variation can be reduced than that in the example of  FIG.  20    by approximately 5 deg. 
       FIG.  18    is a view illustrating a relationship between variation in phase of a load impedance and electric power efficiency according to the second embodiment. The contour diagram of  FIG.  18    illustrates that the electric power efficiency becomes higher at a darker part. Similar to  FIG.  21   ,  FIG.  18    illustrates the peak point Z 10 . From  FIG.  18   , at a point Z 12  (here, 103 deg) of which the phase is shifted by approximately 4 deg with respect to the peak point Z 10 , it is ascertained that the electric power efficiency becomes substantially equivalent (here, approximately ηd=52.1%) to that at the peak point Z 10 . Therefore, it is ascertained that decrease in electric power efficiency with respect to the peak point Z 10  can be curbed to be less than 1% by using the matching circuit  60 . 
     In the foregoing embodiments, embodiments of high frequency amplifiers according to the present disclosure have been described. In the high frequency amplifiers according to the present disclosure, each of the embodiments described above can be arbitrarily changed. 
     For example, the high frequency amplifier  1 A according to the foregoing embodiment includes two of each of the matching circuits  40  and  50 , and the amplifier element portion  10  includes two amplifier elements  11 , but they are not limited to this constitution. The high frequency amplifier  1 A may include the matching circuits  40  and  50  one each or may include three or more matching circuits  40  and  50 . The amplifier element portion  10  may include a single amplifier element  11  or may include three or more amplifier elements  11 . The same also applies to the high frequency amplifier  1 B. Each of the high frequency amplifiers  1 A and  1 B may include the matching circuit  50  or the matching circuit  60  in place of the matching circuit  40 . 
     The number of transistors  13  and the number of bonding wires W 4  are arbitrary, and the numbers thereof may be smaller than eight or may be nine or larger. The numbers of matching circuit patterns  52  and  62  and the numbers of transmission lines  54  and  64  may also be arbitrarily changed in accordance with the number of transistors  13  and the number of bonding wires W 4 . 
     In the foregoing embodiments, the matching circuit pattern  52  include two metal pads  53 , but it is not limited to this constitution. The matching circuit pattern  52  may include only one metal pad  53  or may include three or more metal pads  53 . The matching circuit pattern  52  may include both the metal pads  53  and  63 . The same also applies to the matching circuit pattern  62 . 
     REFERENCE SIGNS LIST 
     
         
           1 A,  1 B High frequency amplifier 
           2  Input terminal 
           3  Output terminal 
           4  Package 
           4   a,    4   b  End wall 
           4   c,    4   d  Side wall 
           4   e  Bottom plate 
           10  Amplifier element portion 
           11  Amplifier element 
           12  Semiconductor substrate 
           13 ,  13 B,  13 C Transistor 
           13 A Transistor (first transistor) 
           13 D Transistor (second transistor) 
           13 E,  13 F,  13 G,  13 H Transistor (third transistor) 
           14  Gate pad 
           15 ,  15 B,  15 C Drain pad 
           15 A Drain pad (first drain pad) 
           15 D Drain pad (second drain pad) 
           15 E,  15 F,  15 G,  15 H Drain pad (third drain pad) 
           16  Source pad 
           20  Branching circuit substrate 
           21  Substrate 
           21   a,    21   b  Long side 
           21   c,    21   d  Short side 
           22  Branching circuit 
           23  Wiring pattern 
           23   a  Metal pad 
           23   b  Film resistor 
           30  Synthetic circuit board 
           31  Substrate 
           31   a,    31   b  Long side 
           31   c,    31   d  Short side 
           32  Synthetic circuit 
           33  Wiring pattern 
           33   a  Metal pad 
           40  Matching circuit 
           50  Matching circuit 
           51  Dielectric substrate 
           52  Matching circuit pattern 
           53  Metal pad 
           53 A Metal pad (first pad) 
           53 B Metal pad (second pad) 
           53   a,    53   b,    53   c,    53   d  Input end 
           53   r,    53   s,    53   u,    53   v  Long side 
           53   t,    53   w  Short side 
           54 ,  54 B,  54 C Transmission line 
           54 A Transmission line (first transmission line) 
           54 D Transmission line (second transmission line) 
           55   a,    55   b,    55   c  Film resistor 
           60 ,  60 X Matching circuit 
           62 ,  62 X Matching circuit pattern 
           63 A Metal pad (first pad) 
           63 B Metal pad (second pad) 
           63 X Metal pad 
           63   a,    63   b,    63   c,    63   d,    63   y  Input end 
           63   r,    63   s,    63   u,    63   v  Long side 
           63   t,    63   w  Short side 
           64 ,  64 B,  64 C Transmission line 
           64 A Transmission line (first transmission line) 
           64 D Transmission line (second transmission line) 
           65   a,    65   b,    65   c  Film resistor 
           80  Metal conductor 
           80 A,  80 B Pattern 
           81 ,  82  Input port 
           83  Output port 
           90  Dielectric substrate 
         C 1  Corner portion (first corner portion) 
         C 2  Corner portion (second corner portion) 
         E Electromagnetic wave 
         D 1 , D 2  Path 
         I Current value 
         L 1 , L 2  Chamfering amount 
         L 11 , L 12 , L 13 , L 14  Width 
         N 1 , N 2 , N 3 , N 4  Virtual straight line (second virtual straight line) 
         N 81 , N 82  Virtual straight line 
         P 1 , P 5  Connection point (first connection point) 
         P 4 , P 8  Connection point (second connection point) 
         P 2 , P 3 , P 6 , P 7  Connection point 
         Q 1 , Q 2 , Q 3 , Q 4  Connection point 
         W 1 , W 2 , W 3 , W 5 , W 6  Bonding wire 
         W 4 , W 42 , W 43  Bonding wire (wire) 
         W 41  Bonding wire (first wire) 
         W 44  Bonding wire (second wire) 
         W 45 , W 46 , W 47 , W 48  Bonding wire (third wire) 
         Z 10  Peak point 
         Z 11 , Z 12  Point 
         ΔZ 1 , ΔZ 2  Phase difference