Patent Publication Number: US-10325584-B2

Title: Active noise cancelling device and method of actively cancelling acoustic noise

Description:
BACKGROUND 
     Technical Field 
     The present disclosure relates to an active noise cancelling device and to a method of actively cancelling acoustic noise. 
     Description of the Related Art 
     As is known, active noise cancelling is becoming more and more used to improve performance of audio systems, such as headphones, headsets, hearing aids, microphones and the like. This trend is also encouraged by recent developments in the field of microelectromechanical systems (MEMS), which provided extremely effective and sensitive devices, such as microphones and speakers, having the additional advantage of very low power consumption. 
     Active noise cancelling essentially consists of detecting acoustic noise produced by noise sources through a microphone at a given location, and using a feedback control based on microphone response to produce acoustic waves that tend to cancel noise by destructive interference in a band of interest (e.g., an audible band roughly comprised between 16 Hz and 16 kHz). 
     Most of known active noise cancelling systems are based on analog circuitry, namely analog filters, because it is normally possible to achieve lower phase delay compared to digital solutions. Filters are in fact included in the feedback control loop and phase delay is well-known to be a critical aspect for stability of feedback system. 
     Apart from a general trend toward digital solutions, analog active noise cancelling systems present some limitations in terms of poor flexibility, accuracy requirements of components, power consumption, area occupation and, in the end, cost. For example, it is quite difficult, or even impossible at all, sometimes, to provide for adjustable filter response and every component, including resistors, should be accurately trimmed to ensure expected performance. Thus, purely analog implementations are not ideally suited to improve miniaturization and flexibility of use. 
     On the other hand, known solutions that involve digital processing based on conventional chains of IIR filters may suffer from low sampling rate typical of audio systems (e.g., 48 kHz) and phase delay, which in turn may undermine stability, as already mentioned. Other active noise cancelling systems envisage higher sampling rates, but these solutions are normally demanding in terms of processing capability. Devices that meet processing requirements (e.g., Digital Signal Processors, DSP) are usually costly and power consuming. 
     BRIEF SUMMARY 
     An aim of the present disclosure is to provide an active noise cancelling device and a method of cancelling acoustic noise that allow some or all of the above described limitations to be overcome and, in particular, favors stability of digital active noise cancelling systems. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       For a better understanding of the disclosure, an embodiment thereof will be now described, purely by way of non-limiting example and with reference to the attached drawings, wherein: 
         FIG. 1  is a block diagram of an audio system including a active noise cancelling device according to an embodiment of the present disclosure; 
         FIG. 2  is a schematic representation of a signal format used in the active noise cancelling device of  FIG. 1 ; 
         FIG. 3  is a more detailed block diagram of a portion of the active noise cancelling device of  FIG. 1 ; 
         FIG. 4  is a detailed block diagram of a first filter of the active noise cancelling device of  FIG. 1 ; 
         FIG. 5  is a detailed block diagram of a second filter of the active noise cancelling device of  FIG. 1 ; 
         FIG. 6  is a block diagram of an audio system including a active noise cancelling device according to another embodiment of the present disclosure; and 
         FIG. 7  is perspective view of a component of the audio system of  FIG. 1 . 
     
    
    
     DETAILED DESCRIPTION 
     In  FIG. 1 , numeral  1  designates an audio system in accordance with an embodiment of the present disclosure and provided with an active noise cancelling function. The audio system  1  comprises a playback unit  2  and a playback unit  3 , both coupled to a signal source  5  that is configured to respectively send audio signals SA 1 , SA 2 . The playback unit  2  and the playback unit  3  may be, for example, left and right earpieces of a headphone assembly. The signal source  5  may be for example, but not limited to, a tuner, a stereo or home theatre system, a cellphone or an audio file player, such as audio file player modules included in a smartphone, a tablet, a laptop or a personal computer. 
     In one embodiment, the audio signals SA 1 , SA 2  supplied by the signal source  5  are oversampled digital signals in single-bit pulse density modulation (PDM) format (e.g., with a sampling frequency of 3 MHz) and the connection to the playback units  2 ,  3  is established through wires  6 . In other embodiments, however, the first audio signals SA 1  and second audio signals SA 2  may be coded in pulse code modulation (PCM) format or may be analog signals. The audio signals SA 1 , SA 2  may represent left audio signals and right channel audio signals, respectively. 
     In the embodiment of  FIG. 1 , the playback unit  2  and the playback unit  3  have the same structure and operation. Accordingly, reference will be made hereinafter to the playback unit  2  for the sake of simplicity. It is however understood that what will be described and illustrated is also applicable to the playback unit  3  and, if provided, to any further playback unit. 
     The playback unit  2  comprises an input interface  7 , a signal processing stage  8 , a microphone  9 , an acoustic noise processing stage  10 , a signal adder  11 , a gain control stage  12 , a D/A stage  13 , an analog amplifier  14  and a loudspeaker  15 , all enclosed within a casing  16 . 
     The input interface  7  is coupled to the signal source for receiving the first audio signal SA 1  and is configured to convert the first audio signal SA 1  into a PDM audio signal SA 1PDM  in single-bit or multibit PDM format. In one embodiment (see  FIG. 2 ), each sample S of a signal in multibit PDM format includes one value bit B V  for the sample value (corresponding to the sample value of single-bit PDM format) and a fixed number N of weight bits B W1 , . . . , B WN  (e.g., five weight bits) defining a sample weight. The input interface  7  may be provided also with wireless communication capability, for receiving audio signals sent by a wireless signal source. 
     The signal processing stage  8  receives the PDM audio signal SA 1PDM  from the input interface  7  and supplies a PCM audio signal SA 1PCM  in PCM format to the signal adder  11 . 
     The signal processing stage  8  includes a set of equalization filters  17  and a processing module  18  with lowpass transfer function and a passband gain which, in one embodiment, may be unity. In one embodiment, the equalization filters  17  may include a cascade of a peak filter  17   a , a notch filter  17   b  and a shelf filter  17   c , as shown in  FIG. 3 . Other sets of filters may be however used, according to the need for specific applications. 
     The output of the equalization filters  17  is a quantized audio signal SA 1QL  in logarithmic multibit PDM format. As herein understood, a logarithmic multibit PDM format is a multibit PDM format in which the weight of each sample is represented in a logarithmic scale. In one embodiment, the weight of each sample is represented in base-2 logarithmic scale. In other words, the weight bits B W1 , . . . , B WN  of each sample represent the base-2 logarithm of the weight of the sample. 
     The processing module  18  applies a gain factor and converts the quantized audio signal SA 1QL  into a PCM audio signal SA 1PCM  in PCM format, which is fed to a first input of the signal adder  11 . In one embodiment, the gain factor may be 1. The lowpass transfer function helps to keep the quantization noise low outside the audio band. 
     The microphone  9  is arranged to detect acoustic noise reaching the inside of the casing  16  from the surrounding environment. In one embodiment, the microphone  9  is a digital microphone and is configured to provide an acoustic noise signal AN PDM  in oversampled PDM format, with the same sampling frequency as the audio signal SA 1  (here 3 MHz). In another embodiment, an assembly including analog microphone and a sigma-delta modulator could be provided in place of the digital microphone. 
     The acoustic noise processing stage  10  receives the acoustic noise signal AN PDM  from the microphone  9  and supplies a filtered audio signal to the signal adder  11 . 
     The acoustic noise processing stage  10  comprises a set of control loop filters  20  and a processing module  21  with lowpass transfer function and passband gain greater than unity. The control loop filters  20  are configured to suppress signal components corresponding to acoustic noise detected by the microphone  9  and may include a cascade of a peak filter  20   a , a notch filter  20   b  and a shelf filter  20   c , as shown in  FIG. 3 . Also in this case, other sets of filters may be used, according to the need for specific applications. 
     The output of the control loop filters  20  is a quantized acoustic noise signal AN QL  in logarithmic multibit PDM format, wherein the weight of each sample is represented in the same logarithmic scale as in the quantized audio signal SA 1QL . 
     The processing module  21  applies a gain factor G 0  (e.g., 100) in the respective passband and converts the quantized acoustic noise signal AN QL  into a PCM acoustic noise signal AN PCM  in PCM format, which is fed to a second input of the signal adder  11 . Also in this case, the lowpass transfer function helps to keep the quantization noise low outside the audio band. 
     The signal adder  11  combines the PCM audio signal SA 1PCM  and the PCM acoustic noise signal AN PCM , respectively received at its first and second input, into a PCM driving signal SD PCM  in PCM format. 
     The gain control stage  12  includes a sigma-delta modulator configured the to convert the PCM driving signal SD PCM  into a PDM driving signal SD PDM  in single-bit or multibit PDM format and to apply a scaling function so that the PDM driving signal SD PDM  complies with the input dynamic of the D/A stage  13 , the analog amplifier  14  and the loudspeaker  15 . 
     The D/A stage  13  includes a lowpass filter and is configured to convert the PDM driving signal SD PDM  into an analog driving signal SD A , which is supplied to the loudspeaker  15  through the amplifier  14 . In one embodiment, the D/A stage  13  may be integrated in the gain control stage  12 , e.g., where a class D amplifier is used. 
     The microphone  9 , the acoustic noise processing stage  10 , the gain control stage  12 , the D/A stage  13 , the analog amplifier  14  and the loudspeaker  15  form an active noise cancelling device  23  that is configured to attenuate acoustic noise within the casing  16  of the playback unit  2 . 
     Acoustic noise is collected by the microphone  9  and converted by the control loop filters  20  into a cancelling component of the driving PDM driving signal SD PDM  that, after further conversion into the analog driving signal SD A , causes the loudspeaker  15  to produce cancelling acoustic wave and suppress acoustic noise by destructive interference. 
     The control loop filters  20  may have any suitable transfer function that effectively achieves noise cancelling and, in one embodiment, they include the peak filter  20   a , the notch filter  20   b  and the shelf filter  20   c , as already mentioned. 
     At least one and, in one embodiment, all of the control loop filters  20  are sigma-delta modulator digital filters, exploiting base-2 logarithmic quantization. 
     The control loop filters  20  may be in the Cascade-of-Integrators FeedBack form (CIFB), which is illustrated by way of example in  FIG. 4  for the peak filter  20   a . However, other sigma-delta modulators, with different structure, could be used. 
     The CIFB peak filter  20   a  comprises a plurality of integrator modules  25 , a plurality of adder modules  26 , a plurality of forward filter modules  27 , a plurality of feedback filter modules  28  and a logarithmic quantizer  30 . 
     The adder modules  26  and the integrator modules  25  are arranged alternated to form a cascade in which each adder module  26  feeds into a respective subsequent integrator module  25  and each integrator module  25  feeds into a respective subsequent adder module  26 . One more adder module  26  is located between the most downstream integrator module  25  and the logarithmic quantizer  30 . 
     Each forward filter module  27  is configured to apply a respective forward filter coefficient W FF1 , W FF2 , . . . , W FFK  to an input signal, i.e., the acoustic noise signal AN PDM  for the peak filter  20   a , and to supply the resulting signal to a first input of a respective one of the adder modules  26 . 
     Each feedback filter module  28  is configured to apply a respective feedback filter coefficient W FB1 , W FB2 , . . . , W FBK-1  to an output signal of the logarithmic quantizer  30  and to supply the resulting signal to a second input of a respective one of the adder modules  26 , except the adder module  26  adjacent to the logarithmic quantizer  30 . 
     In one embodiment, the forward filter coefficient W FF1 , W FF2 , . . . , W FFK  and the feedback filter coefficient W FB1 , W FB2 , . . . , W FBK-1  are programmable and a transfer function of the peak filter  20   a  has a zero at the Nyquist frequency, that improves attenuation of out-of-band quantization noise. 
     In one embodiment, the peak filter  20   a  includes also an internal feedback filter module  31 , that applies an internal feedback filter coefficient to the output of one of the integrator modules  25  and supplies the resulting signal to a third input of one of the upstream adder modules  26 . 
     The logarithmic quantizer  30  quantizes the output signal of the adjacent adder module  26  using a logarithmic scale. In one embodiment, the logarithmic quantizer  30  is a base-2 logarithmic quantizer and provides a multibit PDM signal ranging in module from 2 −M  to 2 M , M being the number of bits for the weight of each sample. 
     The other control loop filters  20  (the notch filter  20   b  and the shelf filter  20   c  in the embodiment described) have the same CIFB structure, possibly with a different number of integrators in the cascade and filter coefficient selected to implement the desired filtering functions. 
     An example of the processing module  21  is illustrated in  FIG. 5  and comprises a gain stage  32  and a plurality of lowpass filter cells  33  in cascade. The gain stage  32  is configured to apply the gain factor G 0  to an input signal of the processing module  21 , i.e., the quantized acoustic noise signal AN QL  received from the control loop filters  20 . The lowpass filter cells  33  in one embodiment are equal to one another and have unity gain. The structure of one of the lowpass filter cells  33  is shown in  FIG. 5 . In one embodiment, the lowpass filter cells  33  comprise each a first gain module  35 , configured to apply a gain factor G 1  to an input signal of the lowpass filter cells  33 ; an adder module  36 ; a delay module  37 ; and a second gain module  38 , configured to apply a gain factor  1 -G 1  to an output signal of the delay module  37 . The adder module  36  combines output signals of the first gain module  35  and of the second gain module  38  and supplies a resulting signal to the delay module  37 , that is configured to apply a unity step delay (i.e., a delay of one sample). 
     In one embodiment, the equalization filters  17  include sigma-delta modulator digital filters in CIFB form. Thus, the equalization filters  17  have the general structure described with reference to  FIG. 4  for the peak filter  20   a , possibly with a different number of integrators and different filter coefficients. However, other sigma-delta modulators, with different structure, could be used. 
     Likewise, the structure of the processing module  18  is similar to the structure of the lowpass amplifier filter  20 , except in that the overall gain is unity and a different number of lowpass filter cells may be included. 
     According to another embodiment, illustrated in  FIG. 6 , an audio system  100  has substantially the structure of the audio system of  FIG. 1  and includes an acoustic sensor  109  in place of the digital MEMS microphone  9 . Moreover, the audio system  100  comprises an additional forward acoustic sensor  130 . 
     The acoustic sensor  109  comprises an analog microphone  109   a  and a sigma-delta A/D converter  109   b  coupled to the microphone  109   a . The sigma-delta A/D converter  109   b  is configured to receive an analog audio signal from the microphone  109   a  and to convert the analog audio signal into the acoustic noise signal AN PDM  in oversampled multibit PDM format. 
     The additional forward acoustic sensor  130  comprises an analog microphone  130   a  and a sigma-delta ND converter  130   b  coupled to the microphone  130   a . The sigma-delta A/D converter  130   b  is configured to receive an analog audio signal from the microphone  130   a  and to convert the analog audio signal into a PDM microphone signal SM PDM  in oversampled PDM format. An input interface  131  of the playback unit  2  receives the PDM microphone signal SM PDM  and converts it into a PCM microphone signal SM PCM , which is then supplied to a third input of the adder module. The input module  131  may include filters and a processing module, similar to the filters and processing modules of the signal processing stage  8  and of the acoustic noise processing stage  10 . 
     A MEMS digital microphone may be used in place of the additional forward acoustic sensor  130  in another embodiment. 
     The solution described above entails several advantages. 
     First, the active noise cancelling function is based on PDM processing and sigma-delta modulator digital filters. On the one side, PDM systems usually exploit a high sampling frequency to produce an oversampled bitstream (3 MHz in the example described). On account of the high sample frequency, latency and delays in the active noise cancelling loop are low, to the benefit of the phase margin, and, accordingly, stability requirements may be easily met. Active noise cancelling function may be thus implemented by reliable fully digital systems. 
     On the other hand, for a given performance level, sigma-delta modulator digital filters have simple structure that is much less demanding in terms of area occupation and power consumption compared to Digital Signal Processors. Thus, also miniaturization is favored to the extent that it is possible to design even in-ear headphones or hearing aids provided with respective active noise cancelling loops and remote processing is not required. For example, see  FIG. 7 , a single package  200  for in-ear headphones may include the MEMS microphone  9  and control circuitry comprising the active noise cancelling device  23 , thus reducing the need for wiring. In this case, the casing  16  is configured to be inserted directly in a user&#39;s ear passage and the package  200  is enclosed within the casing  16  together with the loudspeaker  15 . Also wireless in-ear headphones may be obtained. 
     Multibit PDM coding with a single bit for the sample value and a plurality of bits for the sample weight help to achieve extremely simplified structure. In fact, with this signal format shift registers are enough to implement multipliers, e.g., to apply forward and feedback filter coefficients of the control loop filters. 
     The sigma-delta modulator digital filters are also easily reconfigurable, since it is possible to adjust the forward and feedback filter coefficients by writing registers via software. Therefore, filter trimming is not as critical as with analog solutions. 
     Other advantages are associated with the use of a logarithmic quantizer in the control loop filters, especially a base-2 logarithmic quantizer. Indeed, logarithmic quantizer not only allows a broader dynamic range, but also contributes to reduce quantization noise (out of band noise). Quantization error is in fact correlated to the sample weight, so that the effect on sample having lower absolute value is mitigated. 
     Base-2 quantization puts the sampled signals already in the appropriate multibit PDM format, thereby simplifying processing. 
     Amplification of out-of-band noise present in the PDM signals is avoided by the use of low pass stages in combination with amplification gain. 
     Adding a zero at the Nyquist frequency in at least one of the control loop filters  20  contributes to reduce out-of-band noise and to avoid instability of the structure. 
     The various embodiments described above can be combined to provide further embodiments. All of the U.S. patents, U.S. patent application publications, U.S. patent applications, foreign patents, foreign patent applications and non-patent publications referred to in this specification and/or listed in the Application Data Sheet are incorporated herein by reference, in their entirety. Aspects of the embodiments can be modified, if necessary to employ concepts of the various patents, applications and publications to provide yet further embodiments. 
     These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure.