Patent Publication Number: US-6215451-B1

Title: Dual-band glass-mounted antenna

Description:
FIELD OF THE INVENTION 
     This invention relates to antenna systems for radio-telephone communications, and more particularly, to multiple-band antenna systems usable in cellular and PCS frequency ranges and adapted for coupling through and mounting upon a glass window or other planar dielectric surface. 
     BACKGROUND OF THE INVENTION 
     Recent developments in the wireless telephone communications industry have created the need for wireless subscriber terminals (or “wireless telephones”) capable of operating in two widely displaced frequency ranges. In the United States, the frequency range from approximately 824 to 894 MHz (with some gaps) has been allocated for conventional “cellular” radio telephone service, and the frequency range from approximately 1850 to 1990 MHz has been allocated for a new “Personal Communications System” (PCS) service. Cellular systems, some of which have been in commercial operation since 1984, are relatively mature. Cellular systems provide “blanket” coverage throughout many metropolitan areas and geographically extensive coverage in many other areas where the population density or vehicular traffic are sufficient to warrant coverage. 
     PCS systems, on the other hand, are relatively new, and have a relatively small subscriber base. Some metropolitan areas do not yet have working PCS systems, and even in areas in which one or more PCS systems exist, such systems do not yet provide coverage which is as geographically extensive as that provided by mature cellular systems. As a result, a subscriber to a particular PCS system may often be in a location in which the subscriber&#39;s PCS system is not available, but a cooperative cellular system is available. This could occur, for example, when the subscriber is located within a coverage void in a “home” region generally served by the subscribed PCS system. This could also occur when the subscriber is located outside the home region, such as in a city where the subscriber&#39;s wireless service provider does not operate a PCS system. 
     In order to enable PCS system subscribers to obtain wireless telephone service in areas in which the subscribed PCS system is unavailable, but a cellular system is available, wireless telephone manufacturers have developed wireless telephones capable of operation in both the cellular and PCS frequency bands. For convenient reference, the term “cellular” as applied to frequencies or frequency bands is used herein to refer to the frequency bands allocated in the United States to the Domestic Public Cellular Telecommunications Radio Service (generally, 824 to 894 MHz), and to nearby frequencies, without regard to the type of service, radio protocol standards, or technology actually in use at such frequencies. The term “PCS” as applied to frequencies or frequency bands is used herein to refer to the frequency bands allocated in the United States to Broadband Personal Communications Services (generally, 1850 to 1990 MHz), and to nearby frequencies, without regard to the type of service, radio protocol standards, or technology actually in use at such frequencies. 
     Hand-held wireless telephones are typically equipped with a small, flexible antenna capable of operating, to some extent, in both the cellular and PCS frequency bands. Antennas of this type are very short, compared to the wavelength of the signals to be transmitted and received, and are therefore very inefficient. Such antennas may be adequate when the wireless telephone is used in a location which affords a relatively short, unobstructed RF path to the base station with which communication is desired. However, when the wireless telephone is used in other locations, a better antenna is needed. 
     In particular, when the wireless telephone is used inside a vehicle, the structure of the vehicle both obstructs the RF path between the telephone and the base station, and scatters a substantial amount of the RF energy which would otherwise be transmitted or received by the wireless telephone. Accordingly, it is highly desirable to connect the portable telephone to an efficient antenna located on the exterior of the vehicle. This is especially important when operating in the PCS frequency band. Radio signal propagation characteristics at PCS frequencies are significantly poorer than at cellular frequencies, and the transmitter power allowed at PCS frequencies is significantly lower than the transmitter power allowed at cellular frequencies. 
     A popular type of antenna used in cellular and other vehicular applications is a glass-mounted or window-mounted antenna. Such antennas generally include an external portion semi-permanently affixed to the exterior surface of a vehicle window, and an internal portion semi-permanently affixed to an interior surface of the vehicle window at a position opposite the exterior portion. The interior portion is electrically connected to a suitable transmission line cable which, in turn, may be connected to the mobile telephone transceiver. The internal portion is electrically coupled to the external portion through the glass separating the two portions. The interior portion may incorporate a circuit for matching the impedance of the antenna to the impedance of the transmission line cable and for controlling the impedance of the coupling through the glass. In addition, the interior portion (or an element thereof) may function as a counterpoise. 
     Glass-mounted antennas are preferred in many applications because installing the antenna does not require drilling holes in an exterior vehicle surface for use in mounting the antenna and for passing a transmission line cable. This avoids problems with leakage of air and water into the vehicle, and allows the antenna to be removed from the vehicle without sealing or repairing the holes. Although temporarily installed antennas are available, they are visually obtrusive and require the transmission line cable to be passed through an existing door or window opening. As a result, the transmission line cables are often damaged. 
     A glass-mounted antenna generally as described above, for use at frequencies below those used in cellular and PCS communications, is disclosed in Parfitt U.S. Pat. No. 4,238,799, which is assigned to the assignee of the present application. Glass-mounted antennas for use at cellular frequencies are disclosed in Hadzoglou U.S. Pat. No. 4,839,660, which is assigned to the assignee of the present application, and in Larsen U.S. Pat. No. 4,764,773. It is believed that in each of these antennas, the mechanism by which coupling is achieved through the glass is primarily capacitive. Each of these antennas is designed to operate over a reasonably wide, but nonetheless limited, range of frequencies surrounding an optimum operating frequency. For example, the cellular antennas are disclosed as covering the entire U.S. cellular frequency band. 
     However, none of the antennas described in the aforementioned patents are designed specifically for operation in the PCS frequency band (1850-1900 MHz). Many existing cellular through-the-glass antennas tend to perform poorly in the PCS band due to reasons such as mismatched impedances, poor coupling through the glass, and distorted radiation characteristics. Similarly, many existing PCS antennas tend to perform poorly in the cellular band due to reasons such as mismatched impedances, poor coupling through the glass, and reduced antenna aperture. 
     Although there exist well-known techniques for modifying an existing antenna design to operate at a different frequency, such techniques often cannot be applied when the target operating frequency differs widely from the original operating frequency, because structures and materials may behave electrically in a fundamentally different manner. Moreover, even if the aforementioned antenna designs could be modified to operate at PCS frequencies, the bandwidths of the antennas are not sufficiently wide to allow them to be simultaneously adapted to operate satisfactorily at both cellular and PCS frequencies. Thus, a wireless subscriber using a “dual-band” wireless telephone in a vehicular application would be required to install two separate antennas on the vehicle. 
     Dual-band glass-mounted antennas for use in the 144-148 MHz and 440-450 MHz amateur radio bands have been mentioned in the sales literature of Tandy Corporation of Fort Worth, Tex. (e.g. Radio Shack part number 190-0324), and Larsen Electronics, Inc. of Vancouver, Wash. (e.g. Larsen model number KG 2/70). However, these antennas, and the structures they employ for coupling through the glass and for matching the antenna to the radio transceiver transmission line cable, are not suitable for use in the cellular and PCS frequency bands. 
     In addition, it is believed that these VHF/UHF antenna designs may exploit the serendipitous fact that the higher target operating frequency is almost exactly three times the lower target operating frequency. These antennas generally employ a radiator having upper and lower straight sections separated by a coiled section. The lengths of the straight sections and the parameters of the coiled section are selected such that the total radiator length is equivalent to a half wavelength at VHF. Because of the three-to-one ratio of frequencies, the developed length of the radiator consists of three half-wave sections at UHF. At VHF frequencies, the coil acts as a loading section, with the total radiator acting as a half-wavelength, unity-gain antenna. At UHF frequencies, the coil acts as a phasing element, creating a two element collinear radiator. Thus, this simple configuration works well for the 150 and 450 MHz bands because of the three-to-one ratio of frequencies. 
     This approach to constructing a dual-band antenna cannot be used successfully for the CELLULAR and PCS bands because the ratio of the frequency bands is on the order of two-to-one. The two-to-one frequency ratio tends to transform the low impedances to high impedances, and conversely high impedances to low impedances, between the two bands. This factor complicates the design of a dual-band antenna because it is generally desirable that the antenna present a consistent impedance, approximately matched to the transceiver with which it is to be used, at all operating frequencies. 
     Moreover, existing glass-mounted VHF/UHF dual band antennas employ through-the-glass couplers and associated matching circuitry which are designed to function only with a radiator exhibiting similar base impedances in both frequency bands. Thus, even if the wireless telephone transceiver could tolerate the widely disparate base impedances exhibited by prior-art radiators when used on frequency bands having a two-to-one ratio, these radiators could not be used with prior art through-the-glass couplers. 
     OBJECTS AND SUMMARY OF THE INVENTION 
     It is therefore an object of the present invention to provide a dual-band glass-mounted antenna system for use at disparate frequency bands above 800 MHz. 
     It is another object of the invention to provide a low-loss dual-band through-the-glass coupler for use at disparate frequency bands above 800 MHz. 
     It is a further object of the invention to provide a dual-band antenna element for mounting on a glass or other insulating surface and for use at disparate frequency bands above 800 MHz. 
     It is another object of the invention to provide a dual-band glass-mounted antenna system for use at cellular and PCS frequencies. 
     It is a further object of the invention to provide a low-loss dual-band through-the-glass coupler for use at cellular and PCS frequencies. 
     It is another object of the invention to provide a dual-band antenna element mounting on a glass or other insulating surface and for use at cellular and PCS frequencies. 
     An antenna system constructed according to the present invention comprises a two-element electrical coupling and mechanical attachment unit or coupler to be secured to a planar glass or other dielectric surface, and an antenna element electrically and mechanically coupled to the coupler. The coupler has an internal portion which is secured to an interior surface of the glass, and an external portion which is secured to the exterior surface of the glass at a position opposite the interior portion. The coupler interior portion includes a connection port for electrical connection to a transmission line cable which, in turn, may be connected to the wireless telephone transceiver. 
     The coupler interior portion has a substantially planar conductive sheet element which is oriented in parallel to and secured to the interior surface of the glass. The conductive sheet element incorporates a stepped slot which extends longitudinally to form two transmission line sections. The coupler exterior portion has a similarly shaped planar conductive sheet element which is secured to the exterior surface of the glass in an opposed position. The transmission line sections of the coupler interior portion and coupler exterior portion each operate in the “coupled co-planar strip line mode” on each side of the glass. In addition, the transmission line sections of the coupler interior portion and coupler exterior portion also function cooperatively to form a four-conductor transmission line operating in the “coupled microstrip line odd mode,” thereby achieving coupling through the glass window material. The transmission line characteristics are selected to achieve desired impedances in the cellular and PCS frequency bands. The coupler interior portion and coupler exterior portion thus function in cooperation to provide a through-the-glass coupler which exhibits impedance characteristics within a desired range, and which exhibits minimized insertion loss, at both cellular and PCS frequencies. The coupler interior portion may also act as a counterpoise at some frequencies. 
     The coupler exterior portion has a connection port for electrical connection to the antenna element. The coupler exterior portion preferably includes mechanical supports for the antenna element. The coupler exterior portion may also act as a counterpoise at some frequencies and may have a counterpoise extension section extending a small distance in the transverse direction. The antenna element has a small microstrip matching section which extends, in parallel to the planar sheet, from the coupler exterior portion connection port to the base of a radiating element. The radiating element includes a mechanical support, lower radiator a phasing coil, a middle radiator, a PCS choke assembly, and an upper radiator. The PCS choke assembly exhibits a low impedance at cellular frequencies but a high impedance at PCS frequencies, thereby preventing current from flowing in the upper portion of the radiator at those frequencies. Thus, the entire radiator element acts as a collinear radiator at cellular frequencies, and the lower radiator and middle radiator function as a collinear radiator at PCS frequencies. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other features of this invention will be best understood by reference to the following detailed description of a preferred embodiment of the invention, taken in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a partially exploded perspective view of an antenna system  100  constructed according to the present invention and shown in conjunction with a glass mounting surface with which the antenna system may be used; 
     FIG. 2 is a upward-looking plan view of the interior portion of a through-the-glass coupler for use in the antenna system  100  of FIG. 1; 
     FIG. 3 is a side cross-section view of the interior portion of the through-the-glass coupler of FIG. 2, taken along the view lines  3 — 3  thereof; 
     FIG. 4 is a downward-looking plan view of the exterior portion of the through-the-glass coupler for use in the antenna system  100  of FIG. 1; 
     FIG. 5 is a side cross-section view of the exterior portion of the through-the-glass coupler of FIG. 4, taken along the view lines  5 — 5  thereof; 
     FIG. 6 is a top plan view of a housing and mount for protecting the circuit components of the exterior portion of the coupler and for supporting the radiator element; 
     FIG. 7 is a side cross section view of the housing and mount of FIG. 6, taken along the view lines  7 — 7  thereof; 
     FIG. 8 a  is a Smith chart showing a plot of the input impedance of the through-the-glass coupler of FIGS. 1-6, produced from measurements obtained at the input port of a prototype embodiment of the coupler, with the output port of the coupler connected to a 50 ohm load; 
     FIG. 8 b  is a chart showing a plot of the insertion loss of the through-the-glass coupler of FIGS. 1-6, produced from measurements obtained using a prototype embodiment of the coupler; 
     FIG. 9 is a simplified, upward-looking perspective view of the through-the-glass coupler of FIGS. 1-5, provided to assist in understanding the equivalent circuit of the coupler; 
     FIG. 10 a  is an electrical schematic diagram of a circuit which is electrically equivalent to the through-the-glass coupler of FIGS. 1-5 and  9 , for use in connection with an explanation of the operation of the coupler; 
     FIG. 10 b  is a electrical schematic diagram of a circuit equivalent to the circuit of FIG. 10 a,  but showing the two series impedances of FIG. 10 a  combined to form a traditional pi-network; 
     FIG. 11 is a graph showing the relationship between a parameter {overscore (X)} B , derived from series reactance, and a parameter {overscore (X)} A , derived from the shunt reactances, of the pi-network of FIG. 10, which enable the pi-network to match a selected load impedance; 
     FIG. 12 is a simplified electrical schematic diagram showing an analysis of the operation of the through-the-glass coupler when the input and output ports of the coupler are connected to generators of the same polarity, exciting each of the upper and lower portions of the circuit to operate in the co-planar strip-line mode; 
     FIG. 13 is a simplified electrical schematic diagram showing the current flow in the circuit of FIG. 12, taking into account that in the co-planar strip-line mode the same currents flow on both upper and lower transmission lines; 
     FIG. 14 is a digram showing the distribution of relative voltages on a four-conductor transmission line operating in the coupled microstrip line odd mode; 
     FIG. 15 is a simplified electrical schematic diagram showing an analysis of the operation of the through-the-glass coupler when the input and output ports of the coupler are connected to generators of opposite polarity, exciting the coupled microstrip line odd mode in the four conductors; 
     FIG. 16 is a simplified electrical schematic diagram showing the current flow in the circuit of FIG. 15, taking into account that in the coupled microstrip line odd mode opposite currents flow on both upper and lower transmission lines, and zero-potential points in each circuit allow each side to be analyzed separately; 
     FIG. 17 is a simplified diagram showing the operation of the coupler with a generator connected to the input port and a load impedance connected to the output port; 
     FIG. 18 is a plan view of a simplified transmission line structure to be used to model the behavior of the through-the-glass coupler of FIGS. 1-5 and  9 ; 
     FIG. 19 is a plan view of a portion of the simplified transmission line structure of FIG. 18, illustrating how the behavior of the narrow slot portion thereof may be modeled when operating in the co-planar strip line mode; 
     FIG. 20 is a plan view of a portion of the simplified transmission line structure of FIG. 18, illustrating how the behavior of the wide slot portion thereof may be modeled when operating in the co-planar strip line mode; 
     FIG. 21 is a plan view of a portion of the simplified transmission line structure of FIG. 18, illustrating how the behavior of the narrow slot portion thereof may be modeled when operating in the coupled microstrip line odd mode; 
     FIG. 22 is a plan view of a portion of the simplified transmission line structure of FIG. 18, illustrating how the behavior of the wide slot portion thereof may be modeled when operating in the coupled microstrip line odd mode; 
     FIG. 23 is a schematic diagram of an electrical circuit equivalent to the transmission line structure of FIGS. 18-20, operating in the co-planar strip line mode; 
     FIG. 24 is a schematic diagram of an electrical circuit equivalent to the transmission line structure of FIGS. 18,  21 , and  22 , operating in the coupled microstrip line odd mode; 
     FIG. 25 is a Smith chart plot of the input impedance at selected frequencies of the through-the-glass coupler as modeled according to the transmission line structures and circuits of FIGS. 18-24; 
     FIG. 26 is a table showing the input resistance, input reactance, and voltage standing wave ratio, at selected frequencies, of the through-the-glass coupler as modeled according to the transmission line structures and circuits of FIGS. 18-24; 
     FIG. 27 is a table showing the values of several reactances and the voltage standing wave ratio, at selected frequencies, used in analyzing the through-the-glass coupler as modeled according to the transmission line structures and circuits of FIGS. 18-24, and in particular, the circuits of FIGS. 23 and 24; 
     FIG. 28 is a listing of a computer program which was used to produce the resistance, reactance, and VSWR values tabulated in FIGS. 26 and 27; 
     FIG. 29 is a modified table showing the input resistance, input reactance, and voltage standing wave ratio, at selected frequencies, of the through-the-glass coupler as modeled according to the transmission line structures and circuits of FIGS. 18-24, produced when an alternate value for the electrical length of the wide slot section is employed; 
     FIGS.  30 ( a )-( e ) are exemplary alternate configurations for the transmission lines of the through-the-glass coupler; 
     FIG. 31 is a simplified cross section diagram of the antenna system showing the arrangement of the through-the-glass coupler, the radiator, and a microstrip line section used to match the impedance of the radiator to that of the coupler, taken along view lines  31 — 31  of FIG. 4; 
     FIG. 32 is an electrical schematic diagram of an equivalent circuit including only that portion of the antenna system extending from the output port of the coupler through the radiator; and 
     FIG. 33 is a diagram showing the relative amplitudes and phase of the current distribution along the dual-band antenna/radiator element of FIG. 1, at cellular and PCS frequencies, as determined from current probe measurements. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     preferred embodiment of a dual-band, glass-mounted antenna system  100  constructed according to the present invention is shown generally in FIGS. 1-7. The antenna system  100  comprises a two-element electrical coupling and mechanical attachment unit or coupler  110  to be secured to a planar glass or other dielectric surface or panel  114 , and a dual-band antenna/radiator element  112  which is electrically and mechanically coupled to the coupler  110 . The coupler  110  has a coupler internal portion (“CIP”)  120  which is secured to an interior surface  122  of the glass panel  114 , and a coupler external portion (“CEP”)  116  which is secured to the exterior surface  118  of the glass panel  114  at a position opposite the interior portion  120 . 
     The coupler internal portion  120  has first and second connection points  170  and  166  (FIGS.  1 - 3 ), respectively, forming a first “port” for electrical connection to a suitable transmission line cable  124 . The transmission line cable  124  may be connected to any suitable wireless telephone transceiver (not shown) which operates at cellular frequencies, PCS frequencies, or both. The coupler exterior portion  116  has first and second connection points  270  and  266  (FIGS.  4 - 5 ), respectively, forming a second port for electrical connection to dual-band antenna/radiator element  112 . 
     As will be discussed further in greater detail, the mechanical and electrical structures forming the coupler  110  are engineered to provide an electrical coupling through the glass panel  114  between the first port and the second port. Thus, one skilled in the art will appreciate that the coupler  110  functions as a two-port, reciprocal, electrical network; this observation is useful in understanding the operation and performance of the coupler  110 . 
     Although the dual-band, through-the-glass coupler  110  is discussed herein in the environment of a vehicular application in which the coupler&#39;s first port is connected via a cable to a wireless telephone transceiver and the second port is connected to antenna/radiator element  112 , the coupler  110  may be used advantageously in other applications and with any other generators and users of RF energy. For example, the coupler  110  could be used with an antenna/radiator element other than the radiator  112  disclosed in this application. The coupler  110  could also be used in a stationary application to couple an RF signal source, such as an low-power RF exciter, to an RF signal receiver, such as a power amplifier, located on opposite sides of the glass panel  114 . 
     Moreover, although the coupler  110  is described herein in an application in which the coupling occurs through a glass panel  114 , the coupler  110  could also be used to couple through any other suitable relatively thin dielectric structure, including plastics, Fiberglas, composite materials, and the like, which need not be in a sheet configuration. 
     The preferred embodiment of the coupler  110  disclosed herein is engineered to provide a low-loss, controlled impedance, controlled VSWR coupling between the first and second ports over a first range of frequencies (e.g. 824 to 894 MHz) allocated in the United States to cellular telephone service, and over a second range of frequencies (e.g. 1850 to 1990 MHz) allocated in the United States to PCS communications services. The term “controlled impedance” is used is here to mean that over each of the design frequency ranges, the coupler  110  presents impedance characteristics, which, although they are not constant, do not deviate from a desired impedance by more than an acceptable amount. The term “controlled VSWR” is used here to mean that over each of the design frequency ranges, the coupler  110  presents a VSWR which remains within a desired VSWR range. As a result, the coupler  110  exhibits desirably low insertion loss and VSWR characteristics over the design frequency ranges. The performance of the coupler is discussed further in greater detail. 
     One of skill in the art will appreciate that although a preferred embodiment of the coupler  110  is disclosed herein with mechanical and electrical parameters selected for operation over two particular frequency ranges, the coupler design of the present invention is not limited to those particular frequencies. The mechanical and electrical parameters may be modified, without departing from the basic design of the coupler  110 , to allow the coupler to operate over different frequency ranges, provided that at such frequencies, the structures of the coupler continue to perform the same electrical functions. An analysis of how the coupler functions is discussed further in greater detail. 
     The CIP  120  (FIGS. 1-3) is generally formed as a substantially planar, flexible conductive sheet  154  laminated between an inner insulating sheet layer or film  150  and an outer insulating sheet layer or film  156 . The term “interior” is used here to denote that in a typical vehicular application, CIP  120  is installed on the surface  122  of glass panel  114  facing the vehicle interior. When installed, the inner laminating film layer  150  faces the interior surface  122  of glass panel  114 . The CIP  120  may also act as a counterpoise at some frequencies. 
     The CIP planar conductor  154 , which may act as a counterpoise, may be formed from any suitable flexible, conductive sheet material which is compatible with the inner and outer laminating film layers  150 ,  156 . For example, the CIP planar conductor  154  may be formed from a conductive sheet, foil, or film, such as aluminum, copper, silver, and various conductive alloys, or from a composite material having a conductive component, such as thin printed circuit board material. In addition, the material from which the CIP planar conductor  154  is constructed is preferably environmentally and chemically stable, resistant to corrosion, and is adapted to permit a reliable electrical connection may be made to its surface. In a preferred embodiment of the invention, the CIP planar conductor  154  is formed from brass sheet. The thickness of the CIP planar conductor  154  may range from 0.001 in to 0.050 in. In a preferred embodiment of the invention, the CIP planar conductor  154  is 0.010 in thick. 
     Any suitable insulating film material may be used to form CIP inner and outer laminating film layers  150  and  156 . Layers  150  and  156  provide mechanical support and for the CIP planar conductor  154 , which may be fragile. In addition layers  150  and  156  protect the CIP planar conductor  154  from environmental factors, such as contaminants, which may promote corrosion or deterioration. Because the coupler is intended for use in a vehicular application, they are preferably formed from a material resistant to degradation from strong light, temperature extremes, and typical environmental contaminants, such as water, window cleaner, and the like. However, layers  150  and  156  do not contribute significantly to the electrical performance of the CIP  120 , and therefore, one or more of the layers  150  and  156  may be omitted in some applications. In a preferred embodiment of the invention, the inner and outer laminating film layers  150  and  156  are formed from a polyester film material. The thickness of the inner and outer laminating film layers  150  and  156  may range from 0.005 in to 0.020 in. In a preferred embodiment of the invention, the inner and outer laminating film layers  150  and  156  are 0.005 in thick. 
     Assembly of the CIP inner and outer laminating film layers  150  and  156  and CIP planar conductor  154  into a laminated sheet may be performed by methods well known in the art. As best seen in FIG. 2, the inner and outer laminating film layers  150  and  156  may extend beyond the boundaries of the CIP planar conductor  154  to form an apron  190  of laminating film to provide additional structural support and avoid contamination at the edges of the CIP planar conductor  154 . 
     An adhesive layer  176  is preferably provided on the outside surface of the inner laminating film layer  150  to secure the CIP  120  to the glass panel  114 . Any suitable adhesive which is compatible with the glass panel  114  (or any other dielectric structure to which the coupler is applied), and the CIP inner laminating film layer  150 , may be used. Preferably, the adhesive is a thin pressure sensitive adhesive which may be applied to the inner laminating film layer  150  during manufacture of the coupler, thereby facilitating application of the CIP  120  to the glass panel  114 . For example, a suitable adhesive is available from the 3M Company under the designation SCOTCH VHB 15-mil Foam Tape. However, other adhesives, such as various glues or cement, could also be used. Alternatively, the inner laminating film layer  150  could be thermally bonded to the interior surface  122  of glass panel  114 . 
     The CIP planar conductor  154  is preferably shaped as a generally rectangular sheet having a longitudinally-extending stepped-width slot  160  formed therein. A first electrical connection point  170  is provided on a first side of the slot  160 , and a second electrical connection point  166  is provided on the opposite side of the slot  160 . The electrical connection points  170  and  166  form a first connection port for the coupler  110 . 
     A suitable transmission line cable  124  is preferably electrically and mechanically connected to the CIP planar conductor  154  at the connection points  170  and  166 . For example, as best seen in FIGS. 2-3, a coaxial cable is provided as the transmission line cable  124 . However, other transmission line cables could also be used. The center conductor  128  of the cable is connected to the first connection point  170 , and the outer conductor  126  of the cable is connected to the second connection point  166 . Any suitable means may be used to form these connections, including soldering, spot welding, crimping, and application of conductive paste or glue. A connection point cover  174  is provided to protect the cable and connection points, and may be formed from any suitable insulating material. Relief openings  172 ,  164  are preferably formed in the CIP outer laminating film layer  156  to allow the electrical connection to be made without damaging the layer  156 . 
     The stepped slot  160  divides CIP planar conductor  154  into a first substantially linear conductive strip (including segments  180  and  184 ), second substantially linear conductive strip (including segments  178  and  182 ), which are electrically shorted in the region of  188 . At the operating frequencies of the antenna  100  the conductive strips function as transmission line sections. The dimensions of the strips and the slots, and the dimensions and dielectric constant of adjacent materials, including, especially that of the glass panel  114  or another dielectric to which the coupler is applied and the surrounding air, control the electrical characteristics of the transmission line sections. The conductive strips, in cooperation with complementary conductive strips of the CEP  116 , form a transmission line structure which provides coupling between the connection port of the CIP  120  and the connection port of the CEP  116 . The transmission line structure also provides suitable impedance matching so that the coupler  110  presents desired impedance characteristics at those ports. An analysis of the electrical behavior of the coupler is discussed below in greater detail. 
     In a preferred embodiment of the present invention, designed for dual-bard operation at cellular and PCS frequencies, the CIP planar conductor  154  may be formed as a generally rectangular sheet having an overall length a, and an overall width b. The CIP transmission line slot  160  extends longitudinally from a short end of the sheet and is approximately centered between the long ends of the sheet. The transmission line slot has a wide slot region  158  of width f extending inward a distance e from the short end. 
     The transmission line slot  160  forms first and second transmission line segments  178  and  180 , of approximate width h, in the CIP planar conductor  154 . The CIP transmission line slot  160  also has a narrow slot region  162  of width g extending further inward a distance d from the inner end of the CIP wide slot region  158 . The narrow slot region  162  is not centered. The CIP narrow slot region  162  forms third and fourth transmission line segments  182  and  184  in the CIP planar conductor  154 . The connection points  172  and  166  are intermediately located on transmission line segments  184  and  182  on opposite sides of the CIP narrow slot region  162 . The CIP narrow slot region  162  ends a distance c from the opposite short end of the CIP planar conductor  154 . Thus, in the region designated  188 , the third and fourth transmission line segments  182  and  184  are shorted. In the preferred embodiment, the dimensions are as follows: a=3½ in; b=1.0 in; c=½ in; d=1{fraction (5/16)} in; e=1{fraction (3/8 )} in; f=⅜ in; g={fraction (1/32)} in; and h={fraction (5/16)} in. 
     As best seen in FIGS. 1,  4 , and  5 , the CEP  116  is adapted to be mounted on the exterior surface  118  of glass panel  114 , for supporting a dual-band antenna/radiator element  112  and for providing coupling thereto. Accordingly, although CEP  116  is similar in structure to CIP  120 , CEP  116  incorporates additional structures for mechanically supporting an attached radiator  112 , and for providing impedance matching and electrical connection to the radiator  112 . In addition, CEP  116  may act as a counterpoise for the radiator at some frequencies. If the coupler  110  is used in an application in which the radiator  112  is not present, for example, an application in which transmission line cables are connected to the connection ports of both CIP  120  and CEP  116 , the additional structures may be omitted, and the CEP  116  could be constructed essentially as a mirror image of CIP  120 . The additional structures could, therefore, be considered to be part of the antenna/radiator element  112 . 
     The CEP  116  is generally formed as a substantially planar, flexible conductive sheet  254  laminated between an inner insulating sheet layer or film  250  and an outer insulating sheet layer or film  256 . The term “exterior” is used here to denote that in a typical vehicular application, CEP  116  is installed on the outward-facing surface  118  of glass panel  114 . When installed, the inner laminating film layer  250  faces exterior surface  118  of glass panel  114 . 
     Like the CIP planar conductor  154 , the CEP planar conductor  254  may be formed from any suitable flexible, conductive sheet material which is compatible with the inner and outer laminating film layers  250 ,  256 . For example, the CIP planar conductor  254  may be formed from a conductive sheet, foil, or film, such as aluminum, copper, silver, and various conductive alloys, or from a composite material having a conductive component, such as thin printed circuit board material. The considerations which apply to the selection of CEP planar conductor  254  are essentially the same as those noted for CIP planar conductor  154 . In a preferred embodiment of the invention, the CIP planar conductor  254  is formed from brass sheet. The thickness of the CIP planar conductor  254  may range from 0.001 in to 0.050 in. In a preferred embodiment of the invention, the CIP planar conductor  254  is 0.010 inches thick. 
     Any suitable insulating film material may be used to form CEP inner and outer laminating film layers  250  and  256 . Layers  250  and  256  provide mechanical support and for the CEP planar conductor  254 , which may be fragile. In addition, layers  250  and  256  protect the CEP planar conductor  254  from environmental factors, such as contaminants, which may promote corrosion or deterioration. The considerations which apply to the selection of CEP inner and outer laminating film layers  250  and  256  are essentially the same as those noted for CIP inner and outer laminating film layers  150  and  156 . Because the CEP  116  is intended for use on the exterior of the vehicle, materials are preferably selected to avoid damage from environmental factors. However, layers  250  and  256  do not contribute significantly to the electrical performance of the CEP  116 , and therefore, one or more of the layers  250  and  256  may be omitted in some applications. In a preferred embodiment of the invention, the inner and outer laminating film layers  250  and  256  are formed from polyester film material. The thickness of the inner and outer laminating film layers  250  and  256  may range from 0.005 in to 0.020 in. In a preferred embodiment of the invention, the inner and outer laminating film layers  250  and  256  are 0.005 in thick. 
     Assembly of the CEP inner and outer laminating film layers  250  and  256  and CEP planar conductor  254  into a laminated sheet may be performed by methods well known in the art. As best seen in FIG. 4, the inner and outer laminating film layers  250  and  256  may extend, in certain regions, beyond the boundaries of the CEP planar conductor  254  to form an apron  290  of laminating film to provide additional structural support and avoid contamination at the edges of the CEP planar conductor  254 . 
     An adhesive layer  276  is preferably provided on the outside surface of the inner laminating film layer  250  to secure the CEP  116  to the glass panel  114 . Any suitable adhesive which is compatible with the glass panel  114  (or any other dielectric structure to which the coupler is applied), and the CEP inner laminating film layer  250 , may be used. Preferably, the adhesive is a thin pressure sensitive adhesive which may be applied to the inner laminating film layer  250  during manufacture of the coupler, thereby facilitating application of the CEP  116  to the glass panel  114 . For example, a suitable adhesive is available from the 3M Company under the designation SCOTCH VHB 15-mil Foam Tape. However, other adhesives, such as various glues or cement, could also be used. Alternatively, the inner laminating film layer  250  could be thermally bonded to the exterior surface  118  of glass panel  114 . 
     The CEP planar conductor  254 , which may act as a counterpoise, is preferably shaped as a generally rectangular sheet having a longitudinally-extending stepped-width slot  260  formed therein. A counterpoise extension  292  may be provided to improve impedance matching and radiation characteristics when the coupler  110  is used with radiator  112 . The counterpoise extension  292  may be formed as a rectangular projection extending from one of the long edges of the CEP planar conductor  254 . With the exception of the counterpoise extension  292 , the CEP planar conductor  254  is preferably formed as a mirror-image of the CIP planar conductor  154 . A first electrical connection point  270  is provided on a first side of the slot  260 , and a second electrical connection point  266  is provided on the opposite side of the slot  260 . The electrical connection points  270  and  266  form a second connection port for the coupler  110 . 
     If the coupler  110  is used with radiator  112 , electrical connection therefor may be made at one of the connection points  270  and  266 . In that case, the connection is preferably made at connection  270  to enable the radiator  112  to cooperate with counterpoise extension  292 , which is effectively connected to connection point  266 . If the coupler  110  is used with an antenna/radiator element for which a counterpoise is not desirable, the connection to that element may be made at either connection point  270  or  266 . If the CEP  116  is to be connected to a transmission line cable, the cable may be electrically connected to the CEP planar conductor  254  at the connection points  270  and  266 . Any suitable means may be used to form these connections, including soldering, spot welding, crimping, and application of conductive paste or glue. Relief openings  272 ,  264  are preferably formed in the CEP outer laminating film layer  256  to allow the electrical connection to be made without damaging the layer  256 . 
     Preferably, a coupler exterior circuit housing and radiator mount  130  (FIGS. 1, and  6 - 7 ) is provided to protect the connection points and to mechanically support the radiator  112 . The coupler exterior circuit housing and radiator mount  130  may be formed from any suitable insulating material and prevents forces exerted on the radiator from damaging the CEP  116 , the CEP planar conductor  254 , or the connection points  270 ,  266 . The coupler exterior circuit housing and radiator mount  130  is discussed further in greater detail. 
     The stepped slot  260  divides CEP planar conductor  254  into a first substantially linear conductive strip (including segments  280  and  284 ), and a second substantially linear conductive strip (including segments  278  and  282 , which are electrically shorted in the region of  288 . At the operating frequencies of the antenna  100  the conductive strips function as transmission line sections. The conductive strips, in cooperation with complementary conductive strips of the CIP  120 , form a transmission line structure which provides coupling between the connection port of the CIP  120  and the connection port of the CEP  116 . The transmission line structure also provides suitable impedance matching so that the coupler  110  presents desired impedance characteristics at those ports. An analysis of the electrical behavior of the coupler is discussed further in greater detail. 
     In a preferred embodiment of the present invention, designed for dual-band operation at cellular and PCS frequencies, the CEP planar conductor  254  is formed as a generally rectangular sheet. With the exception of the counterpoise extension  292 , the CEP planar conductor  254  is preferably formed as a mirror-image of the CIP planar conductor  154 , in which components of the CEP planar conductor  254  have the same dimensions as complementary components of CIP planar conductor  154 . Thus, the CEP planar conductor  254  has an overall length a′, and an overall width b′. The CEP transmission line slot  260  extends longitudinally from a short end of the sheet and is approximately centered between the long ends of the sheet. The transmission line slot has a wide slot region  258  of width f′ extending inward a distance e′ from the short end. 
     The transmission line slot  260  forms first and second transmission line segments  278  and  280 , of approximate width h′, in the CEP planar conductor  254 . The CEP transmission line slot  260  also has a narrow slot region  262  of width g′ extending further inward a distance d′ from the inner end of the CEP wide slot region  258 . The narrow slot region  262  is not centered. The CEP narrow slot region  262  forms third and fourth transmission line segments  282  and  284  in the CEP planar conductor  254 . The connection points  272  and  266  are intermediately located on transmission line segments  284  and  282  on opposite sides of the CEP narrow slot region  262 . The CEP narrow slot region  262  ends a distance c′ from the opposite short end of the CEP planar conductor  254 . Thus, in the region designated  288 , the third and fourth transmission line segments  282  and  284  are shorted. 
     The counterpoise extension  292  may be formed as a rectangular member projecting a distance k from one of the long edges and extending along a longitudinal distance l. The counterpoise extension  292  is preferably approximately centered about a transverse line extending through the connection points  270  and  266 . As is known in the art, other shapes and structures could also be used to form a counterpoise extension. In the preferred embodiment, the dimensions are as follows: a′=3½ in; b′=1.0 in; c′=½ in; d′=1{fraction (5/16)} in; e′=1{fraction (3/8 )} in; f′=⅜ in; g′={fraction (1/32)} in; and h′={fraction (5/16)} in; k=¾ in; and l=1¾ in. 
     The CIP  120  and CEP  116  are preferably installed in directly opposing locations on the interior and exterior surfaces  122 ,  118  of glass panel  114 . Considering the CIP  120  in isolation, when excited by a source of RF energy, the CIP conductive strip segments  178 - 182  and  180 - 184  function as transmission line sections operating in the “co-planar strip line mode.” Similarly, considering the CEP  116  in isolation, when excited by a source of RF energy, the CEP conductive strip segments  278 - 282  and  280 - 284  function as transmission line sections operating in the “co-planar strip line mode.” In addition, the conductive strips  178 - 182  and  180 - 184  of the CIP, and  278 - 282  and  280 - 284  of the CEP, function cooperatively to form a four-conductor transmission line operating in the “coupled microstrip line odd mode,” thereby achieving coupling through the material of glass or other dielectric panel  114 . As discussed further in greater detail, the transmission line characteristics are selected to achieve desired impedances in the cellular and PCS frequency bands. The CIP  120  and CEP  116  thus function in cooperation to provide a through-the-glass coupler  110  which exhibits impedance characteristics within a desired range, and which exhibits minimized insertion loss, at both cellular and PCS frequencies. 
     As best seen in FIGS.  1  and  4 - 7 , a suitable dual-band antenna/radiator element  112  is preferably electrically connected to CEP first connection point  270  and mechanically supported by coupler exterior circuit housing and radiator mount  130 . The mount  130  may be formed from any suitable insulating material, such as an insulating plastic. The mount may be formed in a generally rectangular shape having a raised center section  212 , and two lower “wing” portions  208  and  210  adjacent to the center section  212 . The raised portion  212  preferably covers; a cavity or tunnel  216 . An antenna support block  214  is provided to support the radiator element  112  at a position offset in the direction of counterpoise extension  292  from the CEP first connection point  270 . 
     A conductor  202  extends through the tunnel between CEP first connection point  270  and to an antenna attachment stub  204 . The antenna attachment stub  204  is electrically and mechanically connected to a radiator mounting projection  134  extending upward from the antenna support block  214 . An layer of insulating material  206  is provided to maintain a desired separation between the conductor  202  and the underlying CEP planar conductor  254  and counterpoise extension  292 . The conductor  202 , the insulating layer  206 , the CEP planar conductor  254  and the counterpoise extension  292  cooperate to form a transmission line. The transmission line length and characteristic impedance are selected such that the transmission line acts as an impedance matching transformer at PCS frequencies, and complements the low impedance presented by the base of the radiator at cellular frequencies. Operation of the impedance matching transmission line is discussed below in greater detail. 
     As best seen in FIGS. 1, and  6 - 7 , the coupler exterior circuit housing and radiator mount  130  provides a radiator mounting projection  134 . The projection  134  cooperates with a mating mounting projection adapter  192  to form a radiator swivel mount assembly  132 . The mounting projection adapter  192  has a notch for receiving radiator mounting projection  134  and a pivot dowel  186  for retaining the mounting projection  134 . The mounting projection adaptor  192  supports the remaining components of the antenna/radiator element  112 . The radiator swivel mount assembly  132  permits installation of the radiator  112  at an adjustable desired angle, despite variation in the angle of glass panel  114  among various vehicles or other installation sites. 
     The radiator  112  comprises the following electrically relevant components, which are electrically and mechanically connected to one another in this order: a whip adaptor and lower radiator section  136  of length m, comprising mounting projection adapter  192 , and a whip base  194 ; a phasing coil  138  of length n; a middle whip radiator  140  of length p; a PCS band choke assembly  142  of length and an upper whip radiator  148  of length r. In the preferred embodiment, the dimensions are as follows: m=3 in; n=3 in; p=2{fraction (3/4 )}in; q=1 in; and r=3¼ in. 
     The PCS band choke assembly  142  comprises a cylindrical PCS choke sleeve  144  spaced radially from an inner conductor extension of the lower whip radiator  140 . A dielectric filler  146  is provided between the PCS choke sleeve  144  and the inner conductor. The upper end of the PCS choke sleeve  144  is shorted to the center conductor. The PCS band choke assembly  142  forms a shorted transmission line having an effective electrical length of one quarter wavelength at PCS frequencies. The PCS band choke assembly  142  effectively eliminates any current flow beyond the base of the PCS choke sleeve  144  at PCS frequencies. Thus, at PCS frequencies, the radiating section above the phasing coil  138  is approximately one half wavelength. At cellular frequencies, the PCS band choke assembly  142  has little effect, and therefore, the entire assembly above the phasing coil  138  forms a half-wavelength radiator. Other configurations for the PCS choke assembly could also be used. For example, the PCS choke assembly could be implemented using a choke coil which would minimize currents on the upper radiator at PCS frequencies. 
     The lower radiating section of the radiator  112  has an electrical length on the order of one half wavelength at PCS frequencies. Therefore, the base of radiator  112  presents a relatively high impedance, on the order of 500Ω, at PCS frequencies. Thus, the antenna matching section (including conductor  202 ) operates at PCS frequencies to improve the antenna&#39;s VSWR, which would otherwise be undesirably high. In the cellular band, the radiator  112  has an electrical length of approximately one quarter wavelength, and therefore the base of the radiator  112  presents a characteristic impedance on the order of 30-40Ω. At cellular frequencies, the antenna matching section provides a relatively small transformation of the impedance presented by the base of the radiator, resulting in an improved impedance response approaching 50Ω. 
     The phasing coil  138  achieves an “in-phase” condition between the upper and lower co-linear radiators at both cellular and PCS frequency ranges. FIG. 33 is a diagram of the relative amplitudes and phase of the current distribution along the dual-band antenna/radiator element  112  at cellular and PCS frequencies as determined from current probe measurements, using a network analyzer. The current distribution at cellular frequencies is represented by solid line  294 . the current distribution at PCS frequencies is represented by broken line  296 . 
     At cellular frequencies, maximum current occurs at the base  298  of the lower radiator, and at the center of the assembly comprising middle radiator  140 , PCS choke assembly  142 , and upper radiator  148 . The two maximum current regions are “in-phase,” as shown by the direction of the upward-pointing arrows. In the region of the phasing coil  138 , the current is “out-of-phase” with respect to the maximum current regions, as shown by the downward-pointing arrow. Although measurable with a current probe, the current in the region of the phasing coil  138  is effectively non-radiating, and therefore this current does not affect the radiation characteristics of the antenna. Antenna pattern measurements have shown that at cellular frequencies, this radiator configuration exhibits an omnidirectional radiation pattern, with an E-plane beam width in the order of 37°, which is consistent with that expected of a two element collinear array. 
     At PCS frequencies, maximum current occurs at the center of the lower radiator, and at the center of the middle radiator  140  between the top of the phasing coil  138  and the open end of the PCS choke sleeve  144 . The two maximum current regions are “in-phase,” as depicted by the direction of the upward-pointing arrows. In the region of the phasing coil  138 , the current probe measurements show that secondary current peaks occur. Two of the peaks are “out-of-phase” with the primary maximum current regions, while one of the peaks is “in phase.” The symmetry of the secondary current in the region of the phasing coil  138  is believed to be a requirement in order to achieve “in-phase” radiation characteristics for the two-element collinear array formed by dual-band antenna/radiator element  112 . Since the secondary current in the region of the phasing coil  138  is effectively non-radiating, the radiation characteristics of the antenna are not affected. Antenna pattern measurements have shown that at PCS frequencies, this radiator configuration exhibits an omnidirectional radiation pattern, with an E-plane beam width in the order of 31 degrees, which is consistent with that expected of a two-element collinear array. 
     Although not entirely understood, the pitch, number of turns, wire diameter, and coil diameter of the phasing coil  138  seem to be important parameters in achieving proper phasing in both cellular and PCS frequency ranges. 
     The antenna/radiator element  112  described above is one which advantageously provides approximately 2-3 dB of gain over a dipole, or 4-5 dB gain over an isotropic radiator element. However, other types of radiators could also be used. In particular, a simple linear whip radiator of appropriate length may also be used with the coupler  110  to present an impedance equivalent to the radiator  112  described below. For example, a suitable radiator could be constructed in a manner similar to that described for radiator  112 , but omitting the phasing coil and all of the components above it. The resulting radiator is, in essence, a whip radiator having a length of 3.0 in, which is capable of operation in both the cellular and PCS bands. The whip radiator is on the order of a quarter wavelength at cellular frequencies, and on the order of a half wavelength at PCS frequencies. Such a short radiator will exhibit 0 dB gain referenced to a dipole radiator. 
     FIG. 8 a  is a Smith chart showing a plot of the input impedance of the dual-band, through-the-glass coupler  110 . The chart was produced from measurements obtained at the first connection port of a prototype embodiment of the coupler. The second connection port of the coupler was connected to a 50Ω load so that the performance of the coupler could be measured independent of the performance of radiator  112 . The chart shows that the coupler impedance varies from approximately 37 to 42Ω throughout the cellular band, and from approximately 44 to 54Ω throughout the PCS band. In addition, the VSWR remains below 1.5:1 at all frequencies within the cellular and PCS bands. 
     FIG. 8 b  is a chart showing a plot  302  of the insertion loss of the through-the-glass coupler  110  of FIGS. 1-6, at cellular and PCS frequencies, produced from measurements obtained using a prototype embodiment of the coupler. The measured insertion loss of the coupler  110  at cellular frequencies is shown in the region designated  304 . The measured insertion loss of the coupler  110  at PCS frequencies is shown in the region designated  306 . In both frequency bands, the measured insertion loss is on the order of 1.0 dB, indicating that the coupler  110  provides excellent performance. 
     FIG. 9 is a simplified, upward-looking perspective view of the through-the-glass coupler  110  of FIGS. 1-5, which will assist in understanding the equivalent circuit of the coupler. At cellular and PCS frequencies, the conducting strips of the CIP and CEP planar conductors  154 ,  254  behave as transmission lines. It is believed that the fundamental principle on which the coupler works is the excitation of the coupled microstrip line odd mode and the coupled co-planar strip line mode from a signal source on one side (e.g., CEP planar conductor  254 ) of the dielectric  116 , thereby causing these same two modes to exist on the opposite side (e.g., CIP planar conductor  154 ), from which the load impedance (e.g. radiator  112 ) can be driven. The coupled microstrip line odd mode provides the “through the glass” coupling needed in an antenna application. 
     FIG. 10 a  is an electrical schematic diagram of a circuit  402  which is electrically equivalent to the through-the-glass coupler  110  of FIGS. 1-5 and  9 . The circuit  402  represents a reciprocal two-port network, in which a first port corresponds to CIP first connection point  170  and CIP second connection point  166 , and a second port corresponds to CEP first connection point  270  and CEP second connection point  266 . Any reciprocal two-port network can be represented by a pi-network. FIG. 10 b  is a electrical schematic diagram of a circuit  404  which is equivalent to the circuit of FIG. 10 a,  but in which the two series impedances Z B  of FIG. 10 a  are combined to form a traditional pi-network. The properties of the pi-matching network are well known. A load resistance R L  can be matched to a generator with an internal resistance R g =R L  by a range of values for X A  and X B . The requirement is that X A  and X B  be of the opposite sign and that            X   B       R   L       =         X   _     B     =       -                    X   A       R   L         1   +       (       X   A       R   L       )     2           =     -                      X   _     A       1   +       X   _     A   2         .                           
     The parameter tolerances are less if X A  is of the order of R L  or larger and X B  is then smaller. 
     FIG. 11 is a graph  406  showing the relationship  408  between a parameter {overscore (X)} B , derived from series reactance, and a parameter {overscore (X)} A , derived from the shunt reactances, of the pi-network  404  which enable the pi-network to match a selected load impedance. The maximum value for {overscore (X)} B  is −{overscore (X)} A . 
     The availability of a range of values for X A  and X B  simplifies the design of coupler  110 . The coupler may be constructed by selecting transmission line characteristics such that the transmission line network provides a suitable pair of values for X A  and X B  at each desired operating frequency. Values for X A  in the range of R L  and larger are suitable. The pi-network can also match complex load impedances, as is well known. In practice, small values of X A  and X B  work poorly because circuit losses make the impedance match poor. In the following discussion,            Y   A     =     1     j                   X   A           ,                  and                   Y   B       =       1     j                   X   B         .                       
     By using even and odd excitation at the input and output ports in the equivalent circuit shown in FIG. 10, and correspondingly, at the two ports of the transmission line coupler, the input currents can be compared to establish the values of the reactances X A  and X B  in the equivalent circuit, in terms of reactances evaluated for the transmission line circuits. Even excitation is obtained by connecting voltage generators of the same polarity at each port, while odd excitation is obtained by connecting voltage generators of opposite polarity at the two ports. 
     FIG. 12 is a simplified electrical schematic diagram showing an analysis of the operation of the through-the-glass coupler  110  when the input port (connection points  270 ,  266 ) and output port (connection points  170 ,  166 ) of the coupler are connected to generators of the same polarity. This configuration excites each of the upper and lower portions  410 ,  412  of the circuit (equivalent to CEP planar conductor  254  and CIP planar conductor  154 ) to operate in the co-planar strip-line mode. These two co-planar strip line modes interact to form a coupled co-planar strip line mode (mode a). Let both upper and lower lines be driven by voltage generators with voltage 2V g . The currents I a1  and I a2  that flow are given by I a1 =2V g Y a1  and I a2 =2V g Y a2  where Y a1  and Y a2  are the two input admittances looking toward Z 1  and Z 2  respectively on the transmission line. This drive excites the coupled co-planar strip line mode on both the upper and lower transmission lines  410  and  412 . FIG. 13 is a simplified electrical schematic diagram  414  showing the current flow in the circuit of FIG.  12 . The same currents flow on both upper and lower transmission lines  410  and  412 . The generator supplies current I g1 =I a1 +I a2 =2V g (Y a1 +Y a2 )=2V g Y a . The same excitation applied to the equivalent circuit in FIG. 10 would result in the same currents. There will be zero current in the impedance Z B  and a current 2V g Y A  in Z A , and thus the impedance Z A  in FIG. 10 can be equivalently identified as 1/Y a . 
     FIG. 14 is a diagram showing the distribution of relative voltages on a four-conductor transmission line, formed by CEP and CIP planar conductors  254 ,  154  operating in the coupled microstrip line odd mode. For the coupled microstrip odd mode, planes AB and CD are zero potential or “virtual” short circuits. 
     FIG. 15 is a simplified electrical schematic diagram showing an analysis of the operation of the through-the-glass coupler  110  when the input port (connection points  270 ,  266 ) and output port (connection points  170 ,  166 ) of the coupler of the coupler are connected to generators of opposite polarity. With the two generators connected with opposite polarities, the coupled microstrip line odd mode (mode b) is excited. Z 1  and Z 2  are split in two to show the zero-potential points, P 1  and P 2 . When Z 1  is a short circuit and mode b is excited, a virtual short circuit is seen at P 1 . When Z 2  is an open circuit and mode b is excited, an open circuit is also seen at P 2 . Points  1 - 2  and  3 - 4  (connection points  266 - 270 , and  166 - 170 , respectively) have a potential difference 2V g . Likewise points  1 - 3  and  2 - 4  have a potential difference of 2V g , and hence the coupled microstrip line odd mode must exist on the structure. 
     FIG. 16 is a simplified electrical schematic diagram  420  showing the current flow in the circuit of FIG.  15 . The simplified analysis exploits the facts that in the coupled microstrip line odd mode, opposite currents flow on both upper and lower transmission lines  416  and  418 , and zero-potential points in each circuit allow each side to be analyzed separately. The coupled microstrip odd mode currents is are given by I b1 =2V g Y b1  and I b2 =2V g Y b2 , where Y b1  and Y b2  are the input admittances due to the coupled microstrip odd mode. The total current is I b =I b1 +I b2 =2V g Y b . When odd excitation is applied to the equivalent circuit in FIG. 10, the input current is readily found to be given by 2V g (Y A +Y B ) and must equal 2V g Y b . From this relation, and the previous one (Y A =Y a ), it is found that          Y   B     =         Y   b     -       Y   a                   and                   Z   B         =       1       Y   b     -     Y   a         .                       
     By superimposing the solutions for mode a (FIGS. 12-13) and mode b (FIGS.  14 - 16 ), the equivalent circuit  422  of FIG. 17 is obtained. FIG. 17 is a simplified diagram showing the operation of the coupler with a generator connected to the input port and a zero impedance load (short circuit) connected to the output port. The input is driven by a generator with voltage 4V g . The output has zero voltage across the slot and short circuit current (I b )−(I a )=2V g (Y b −Y a ) where Y b =Y b1 +Y b2  and Y a =Y a1 +Y a2 . In the equivalent circuit in FIG. 10, the output short circuit current is 2V g Y B  and equals 2V g (Y b −Y a ). The admittance Y a  is associated with the coupled co-planar strip line mode. The admittance Y b  is associated with the coupled microstrip line odd mode. When the structure has some asymmetry and is driven in an unbalanced way, it is also possible to excite both the coupled microstrip line even mode and the antenna mode, but neither of these two modes have an electric field and voltage across the gap between strips on the input and output sides. Accordingly, neither of these two modes contribute to coupling through the glass  114  or another dielectric material to which the coupler may be directed. 
     When a finite load impedance is connected across the output terminals, with a generator on the input side, the only difference in the circuit operation is that the amplitudes of the coupled co-planar strip line mode and the coupled microstrip line odd mode will no longer be equal. The analysis carried out has identified how to find the equivalent circuit parameters Z A  and Z B  in the circuit representing the coupler (see FIGS. 10 a - 10   b ). Z A  and Z B  are given by          Z   A     =     1     Y   a                       
     and          Z   B     =       1       Y   b     -     Y   a                    .                     
     FIG. 18 is a plan view of a simplified transmission line structure  424  to be used to model the behavior of the coupler  110  of present invention. The admittance Y a  must be replaced by the transmission line admittance seen across the gap. The admittance Y b  must be replaced by the transmission line admittance seen between conductors on the opposite sides of the dielectric. These admittances depend on the characteristics of the transmission lines and how they are terminated. The width and spacing of the conducting strips, the termination of the transmission lines on each side in open circuits or short circuits, and the lengths of the transmission lines, are chosen so as to obtain impedance matching between the signal source and the load impedance (antenna) at two (or more) different frequencies. 
     The structure  424  which is analyzed in the following discussion is the same as the coupler  110  of FIGS. 1-5, with the exception that the narrow slot  426  is centered about the longitudinal axis of the transmission line. This modification simplifies the analysis, but it is believed that the change has a negligible effect on the results. The analysis of this simplified model was undertaken to verify that the principles of operation described above apply to the dual-band coupler  110 . The glass thickness is taken as {fraction (5/32)} in, and the dielectric constant is taken as 7. Because the electric field is partly in air and partly in the dielectric, the effective dielectric constants range between 1 and 7, depending on the mode and the transmission line segment being considered. The effective dielectric constants, mode characteristic impedances, and effective wavelengths were found using well known formulas for microwave lines. 
     The electrical length of a line in radians is 2πl/λ e , where l is the physical length and λ e  is the effective wavelength equal to the free-space wavelength λ 0  (13.12 in at 900 MHz) divided by the square root of the effective dielectric constant, {square root over (ε e +L )}. End effects may be estimated and used to modify the line length somewhat from their physical length. 
     FIGS. 19 and 20 are directed to modeling the behavior of the coupler transmission lines in the coupled co-planer strip line mode, in which the upper and lower transmission lines are analyzed in parallel. FIG. 19 is a plan view of a portion  432  of the simplified transmission line structure  424  of FIG. 18, illustrating how the behavior of the narrow slot portion thereof  426  may be modeled when operating in the coupled co-planar strip line mode. The following calculated values may be used to model the structure: ε e =3.07; λ e =7.49 in (900 MHz); and Z c =91Ω. l=1 in corresponds to 480 or 0.84 radians. The narrow slot portion  426  appears longer because current flows around the corner  428 , following a longer path. Accordingly, the length l is preferably increased by an estimated 0.12 in, to give an electrical length of 0.94 radians at 900 MHz. 
     FIG. 20 is a plan view of a portion  434  of the simplified transmission line structure  424  of FIG. 18, illustrating how the behavior of the wide slot portion thereof  430  may be modeled when operating in the co-planar strip line mode. The following calculated values may be used to model the structure: ε e =2.15; λ e =8.95 in (900 MHz); and Z c =280Ω. Fringing capacitance  490  (C f ) of 0.22 pF (−j800Ω in shunt at 900 MHz) is preferably added at the step junction  488 . The electrical length of the 1⅜ in line is 0.965 radians at 900 MHz. 
     FIGS. 21 and 22 are directed to modeling the behavior of the coupler transmission lines in the coupled microstrip line odd mode. FIG. 21 is a plan view of a portion  436  of the simplified transmission line structure  424  of FIG.  18 , illustrating how the behavior of the narrow slot portion  426  thereof may be modeled when operating in the coupled microstrip line odd mode. The following calculated values may be used to model the structure: ε e =5.22; λ e =5.74 in (900 MHz); and Z c =31Ω. l=1 in corresponds to an electrical length of 62.7° or 1.095 radians. This length is preferably increased by an estimated 0.37 in, since current in this mode will spread further beyond the shorted end  428  of the narrow slot  426 , resulting in an electrical length of 1.5 radians at 900 MHz. A fringing capacitance  440  (C f ) of 0.22 pF is preferably added at the step junction  488 . 
     FIG. 22 is a plan view of a portion  438  of the simplified transmission line structure  424  of FIG. 18, illustrating how the behavior of the wide slot portion thereof  430  may be modeled when operating in the coupled microstrip line odd mode. The following calculated values may be used to model the structure: ε e =5.5; λ e =5.59 in (900 MHz); and Z c =49Ω. The 1⅜ in line has an electrical length of 88.5° or 1.54 radians. This length is preferably increased to 1.64 radians to account for the open-circuit end capacitance  442  (C oc ) between the upper and lower strips. 
     FIG. 23 is a schematic diagram of an electrical circuit  444  equivalent to the transmission line structure  424  of FIGS. 18-20, operating in the co-planar strip line mode. Circuit  444  is used to derive the following:            j                   X   1       =     1     Y   a1         ;       j                   X   4       =     1     Y   a2         ;                   
     and          j                   X   a       =       1       Y   a1     +     Y   a2         =     j                     X   A     .                         
     FIG. 24 is a schematic diagram of an electrical circuit  446  equivalent to the transmission line structure  424  of FIGS. 18,  21 , and  22 , operating in the coupled microstrip line odd mode. Circuit  446  is used to derive the following:            j                   X   B        1     =     1     Y   b1         ;       j                   X   B        5     =     1     Y   b2         ;       j                   X   b       =     1       Y   b1     +     Y   b2           ;       j                   X   B       =       1       Y   b     -     Y   a         =     j                         X   a          X   b           X   a     -     X   b         .                           
     With the analysis described above, the performance of the model of the dual-band coupler  110  was found to be substantially in accord with a prototype embodiment of the dual band coupler  110  described herein, with the exception that the model predicted a VSWR somewhat greater than 1.5 in the PCS frequency band and a smaller insertion loss than that measured in the frequency range between the two bands. The input-normalized resistance and reactance and the VSWR, which were predicted by the model, are shown graphically in FIG.  25  and in tabular form in FIGS. 26-27 and  29 . The model analyzed neglected the offset of the narrow slot from the longitudinal axis, and did not take into account the laminations or the dielectric loss in the glass. Estimated values were used for the relative dielectric constant of the glass, and the end effect for the transmission lines. The formulas used to calculate the characteristic impedances and the effective dielectric constants for the various transmission line modes are believed to be accurate to within a few percent, but even small inaccuracies would account for some difference between the calculated and measured results. 
     In order to harmonize the calculated results with the measured results, the model which was quantitatively analyzed to produce the results of FIGS. 25-27 incorporated one change to a parameter derived from the physical dimensions of the coupler. This change was the reduction of the electrical length of the transmission line representing the coupled co-planar strip line mode in the wide slot region from 0.965 radians (as calculated from physical dimensions; see FIG.  20 ) to 0.9 radians at 900 MHz. This one change brought the calculated VSWR in both the cellular and PCS frequency bands to the desired value of 1.5:1 or less. The predicted performance of a dual-band coupler  110 , is shown in FIG.  29 . 
     FIG. 25 is a Smith chart  448  showing a plot  450  of the predicted input impedance of the through-the-glass coupler, at selected frequencies. Circle  452  corresponds to a voltage standing wave ratio (VSWR) of 1.5:1; the region within the circle  452  corresponds to VSWRs below 1.5:1, representing an acceptable impedance match. The Smith chart plot  450  remains near the origin of the chart, and within the circle  450 , throughout the 824 to 894 MHz cellular frequency band. At frequencies above the cellular frequency band but below the PCS frequency band the plot deviates widely from the origin, indicating that the impedance match is poor at those frequencies. Within the 1850 to 1990 MHz PCS frequency band, the Smith chart plot  450  returns to the region near the origin of the chart and within the circle  450 . Thus, in both the cellular and PCS frequency bands, the performance of the coupler, as predicted by the model described above, is excellent. 
     At frequencies approximately midway between the cellular and PCS bands, the VSWR reaches a peak value of 54.89, which corresponds to an insertion loss; of 11.5 dB. The measured insertion loss shown in FIG. 8 b  was somewhat larger than 14 dB. Overall, the agreement between measured and predicted values is considered to be very good, in view of the limitations involved in the analyzed model. In particular, it is believed that circuit losses, which were neglected in the model, would have increased the predicted insertion loss had their effects been included in the calculations. 
     FIG. 26 is a table  454  showing the calculated input resistance, input reactance, and voltage standing wave ratio (VSWR), at selected frequencies, of the through-the-glass coupler, according to the model described above. FIG. 27 is a table  456  showing key reactances X A  and X B , and corresponding VSWRs, used in analysis of the circuits of FIGS. 23 and 24 at selected frequencies. FIG. 28 is a listing  460  of a computer program for calculating the resistance, reactance, and VSWR values tabulated in FIGS. 26 and 27. The table of FIG. 27 shows that the obtained pairs of values for the reactances X A  and X B  provide good impedance matching in both the cellular and PCS frequency bands. It is believed that the analysis presented has established the operating principles of the dual-band coupler  110 , and the calculated numerical results have verified the correctness of those operating principles. 
     The predicted performance of a dual-band coupler  110 , including the input-normalized resistance and reactance and the VSWR, calculated according to the model, and using a wide slot region transmission line length of 0.965 radians, is shown in FIG.  29 . 
     FIGS.  30 ( a )-( e ) are exemplary alternate configurations for the transmission lines of the through-the-glass coupler. The transmission lines and the way they are terminated must be designed so as to obtain a suitable pair of values for X A  and X B  to provide impedance matching at two or more frequencies. Steps in characteristic impedance (such as step changes in conductor widths or spacing) will effect how the transmission line transforms an open circuit or short circuit impedance into an X A  or X B  as a function of frequency. As best seen in FIG.  30 ( a ), the transmission lines  482   a  may employ a single impedance step. As best seen in FIG.  30 ( b ), the transmission lines  482   b  could also employ two impedance steps. As best seen in FIG.  30 ( c ), the transmission lines  482   c  could also employ shunt capacitive loading in the form of a short stub. Alternatively, as best seen in FIG.  30 ( d ), the transmission lines  482   d  could employ shunt inductive loading in the form of notches. As best seen in FIG.  30 ( e ), the transmission lines  482   e  could also employ a taper. Moreover, the taper may be combined with a step. Such modifications or alternatives may be employed to alter the impedance transforming properties of the transmission lines. These and other microstrip techniques are well known in the microwave field. One of ordinary skill will appreciate how such modifications of the coupler  110  may be applied consistent with the spirit of the present invention. 
     FIG. 31 is a simplified cross section diagram  484  of the antenna system showing the arrangement of the through-the-glass coupler  110 , the radiator  112 , and a microstrip line section (including conductor  202 ) used to match the impedance of the radiator  112  to that of the coupler  110 . FIG. 32 is an electrical schematic diagram of an equivalent circuit  486  including only that portion of the antenna system extending from the output port of the coupler  110  through the radiator. A broken vertical line designated  218  bisects CEP narrow slot region  262  in an area  492  near CEP first and second connection points  270  and  266 . The electric field across the slot at  492  acts as a series voltage source V g    494  to drive the antenna transmission line, which comprises a single conductor  202  over the conducting segments  282  and  284 . Segments  282  and  284  act as a ground plane. The transmission line conductor  202  is connected to conducting segment  284 , and therefore to the ground plane, at the connection point  270 . Accordingly, the portion  222  of conductor  202  to the left of line  218 , and the portion  224  of conducting segment  284  to the left of line  218 , form a shorted transmission line section. The relatively short length of the shorted transmission line section behaves as an inductor in series with the equivalent voltage source  494  represented by the electric field across the narrow slot. This series inductance is shown in the equivalent circuit of FIG. 32 as inductor  220 . This inductance is relatively unimportant at cellular frequencies but becomes a more significant part of the antenna matching network at PCS frequencies. 
     The above-described embodiments of the invention are merely examples of ways in which the invention may be carried out. Other ways may also be possible, and are within the scope of the following claims defining the invention.