Patent Publication Number: US-2011049997-A1

Title: Wireless power feeder and wireless power transmission system

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a wireless power feeder for feeding power by wireless and a wireless power transmission system. 
     2. Description of Related Art 
     A wireless power feeding technique of feeding power without a power cord is now attracting attention. The current wireless power feeding technique is roughly divided into three: (A) type utilizing electromagnetic induction (for short range); (B) type utilizing radio wave (for long range); and (C) type utilizing resonance phenomenon of magnetic field (for intermediate range). 
     The type (A) utilizing electromagnetic induction has generally been employed in familiar home appliances such as an electric shaver; however, it can be effective only in a short range of several centimeters. The type (B) utilizing radio wave is available in a long range; however, it cannot feed big electric power. The type (C) utilizing resonance phenomenon is a comparatively new technique and is of particular interest because of its high power transmission efficiency even in an intermediate range of about several meters. For example, a plan is being studied in which a receiving coil is buried in a lower portion of an EV (Electric Vehicle) so as to feed power from a feeding coil in the ground in a non-contact manner. Hereinafter, the type (C) is referred to as “magnetic field resonance type”. 
     The magnetic field resonance type is based on a theory published by Massachusetts Institute of Technology in 2006 (refer to Patent Document 1). In Patent Document 1, four coils are prepared. The four coils are referred to as “exciting coil”, “feeding coil”, “receiving coil”, and “loading coil” in the order starting from the feeding side. The exciting coil and feeding coil closely face each other for electromagnetic coupling. Similarly, the receiving coil and loading coil closely face each other for electromagnetic coupling. The distance (intermediate distance) between the feeding coil and receiving coil is larger than the distance between the exciting coil and feeding coil and distance between the receiving coil and loading coil. This system aims to feed power from the feeding coil to receiving coil. 
     When AC power is fed to the exciting coil, current also flows in the feeding coil according to the principle of electromagnetic induction. When the feeding coil generates a magnetic field to cause the feeding coil and receiving coil to magnetically resonate, large current flows in the receiving coil. At this time, current also flows in the loading coil according to the principle of electromagnetic induction, and power is taken out from a load R connected in series to the loading coil. By utilizing the magnetic field resonance phenomenon, high power transmission efficiency can be achieved even if the feeding coil and receiving coil are largely spaced from each other. 
     CITATION LIST 
     Patent Document 
     [Patent Document 1] U.S. Pat. Appln. Publication No. 2008/0278264
 
[Patent Document 2] Jpn. Pat. Appln. Laid-Open No. 2006-230032
 
[Patent Document 3] International Publication Pamphlet No. WO2006/022365
 
[Patent Document 4] U.S. Pat. Appln. Publication No. 2009/0072629
 
     In order to generate the magnetic field resonance phenomenon, the drive frequency of a power circuit needs to be made to coincide with the resonance frequency when AC power is fed to the exciting coil and feeding coil. For example, Patent Document 2 discloses a technique of detecting whether the drive frequency and resonance frequency coincide with each other. More specifically, in Patent Document 2, the voltage phase of a primary coil L 1  corresponding to the feeding coil is compared to a reference phase to thereby detect whether a resonance state exists or not (refer to paragraphs [0043] and [0044] and FIG. 1 of Patent Document 2). However, in the case of Patent Document 2, the voltage waveform itself of the primary coil L 1  to be resonated is measured, the resonance characteristics (Q-value) is easily be degraded by the measurement procedure. In other words, a system configuration disclosed in patent Document 2 is susceptible to the measurement procedure. 
     SUMMARY 
     The present invention has been made in view of the above problem, and a main object thereof is to detect the phase of feed power while suppressing influence on the resonance characteristic in a wireless power feeding technique of a magnetic resonance type. 
     A wireless power feeder according to a first aspect of the present invention feeds power by wireless from a feeding coil to a receiving coil at the resonance frequency of the feeding coil and receiving coil. The wireless power feeder includes: a power circuit; a feeding coil; an exciting coil that is magnetically coupled to the feeding coil and feeds AC power fed from the power circuit to the feeding coil; and a phase detection circuit that detects the phase difference between the voltage phase and current phase of the AC power fed from the power circuit. The power circuit includes first and second current paths and makes first and second switches connected in series respectively to the first and second current paths alternately conductive to feed the AC power to the exciting coil. The phase detection circuit measures the phase of current passing through both or one of the first and second switches to achieve measurement of the current phase of the AC power. 
     A wireless power feeder according to a second aspect of the present invention also feeds power by wireless from a feeding coil to a receiving coil at the resonance frequency of the feeding coil and receiving coil. The wireless power feeder includes: a power circuit that feeds AC power to the feeding coil at the drive frequency; a feeding coil circuit that includes the feeding coil and a capacitor and resonate at the resonance frequency; and a phase detection circuit that detects the phase difference between the voltage phase and current phase of the AC power fed from the power circuit. The power circuit includes first and second current paths and makes first and second switches connected in series respectively to the first and second current paths alternately conductive to feed the AC power to the feeding coil. The phase detection circuit measures the phase of current passing through both or one of the first and second switches to achieve measurement of the current phase of the AC power. 
     The use of the power circuit that operates as a switching power source for the feeding coil can enhance the efficiency of power transmission from the power circuit to feeding coil. When the drive frequency of the power circuit and resonance frequency are made coincide with each other, the power transmission efficiency in the entire system can be enhanced. The current phase is measured from current passing through a switch included in the power circuit, so that a measurement load is not directly applied to the feeding coil. Thus, it is possible to monitor whether a resonance state is maintained by detecting the phase difference between the voltage phase and current phase while suppressing influence on the resonance characteristics of the feeding coil. 
     The wireless power feeder may further include a drive frequency tracking circuit that adjusts the drive frequency of the power circuit so as to reduce the detected phase difference to allow the drive frequency to track the resonance frequency. In this case, it is possible to allow the drive frequency to track the resonance frequency, thereby making it easy to maintain high power transmission efficiency. 
     The first and second switches may each be a field-effect transistor. In this case, the phase detection circuit may measure the current phase from a change in voltage applied to first resistors that are connected in series between the source and ground of the first switch and between the source and ground of the second switch. Further, the phase detection circuit may measure the voltage phase from a change in the source-drain voltage of both or one of the first and second switches. The phase detection circuit may measure the voltage phase from a change in intermediate potential taken from the middle of second resistors that are connected in parallel to the source-drain of both or one of the first and second switches. 
     The wireless power feeder may further include: a first waveform rectifier that converts an analog waveform having the same phase as that of a current waveform of the AC power into a digital waveform; and a second waveform rectifier that converts an analog waveform having the same phase as that of a voltage waveform of the AC power into a digital waveform. The phase detection circuit may compare the edges of two digital waveforms to detect the phase difference. The digitization makes clear the reference point used for comparing a current waveform and a voltage waveform, making it easy for the phase detection circuit to identify the phase difference. 
     A wireless power feeder according to a third aspect of the present invention also feeds power by wireless from a feeding coil to a receiving coil at the resonance frequency of the feeding coil and receiving coil. The wireless power feeder includes: a resonance circuit that includes a first coil and a capacitor which are connected in series; a first switch that controls supply of power fed from a first direction to the resonance circuit; a second switch that controls supply of power fed from a second direction to the resonance circuit; a power transmission control circuit that makes the first and second switches alternately conductive to cause the resonance circuit to resonate to transmit AC power from the first coil serving as the feeding coil to the receiving coil; a second coil that generates inductive current using a magnetic field generated by the AC power; and a phase detection circuit that detects the phase difference between the voltage phase and current phase of the AC power. The phase detection circuit measures the phase of the inductive current flowing in the second coil to achieve measurement of the current phase of the AC current. 
     This wireless power feeder can directly drive the feeding coil without use of the exciting coil. This contributes to a reduction in the manufacturing cost and size of the wireless power feeder. When the drive frequency of the power circuit and resonance frequency are made coincide with each other, the power transmission efficiency in the entire system can be enhanced. Inductive current is made to occur in the second coil (detection coil) by means of a magnetic field generated by AC current, and the current phase is measured from the inductive current, so that a measurement load is not directly applied to the feeding coil. Thus, it is possible to monitor whether a resonance state is maintained by detecting the phase difference (deviation) between the voltage phase and current phase while suppressing influence on the resonance characteristics of the feeding coil. 
     A current path passing through the first and second switches and a current path passing through the resonance circuit may be separated by a coupling transformer. AC power may be fed to the resonance circuit through the coupling transformer. 
     The wireless power feeder may further include a drive frequency tracking circuit that adjusts the drive frequency of the power transmission control circuit so as to reduce the detected phase difference to allow the drive frequency to track the resonance frequency. In this case, it is possible to allow the drive frequency to track the resonance frequency, thereby making it easy to maintain high power transmission efficiency. 
     The power transmission control circuit may make a coil in the resonance circuit operate not as the feeding coil but as the exciting coil so as to feed power to another coil serving as a feeding coil. 
     The second coil may be wounded around a toroidal core. A part of the first coil may be made to pass through the toroidal core to constitute a coupling transformer by the first and second coils. By sharing the toroidal core between the first and second coils, it is possible to allow the second coil to satisfactorily generate the inductive current. 
     A resistor may be connected in parallel to both ends of the second coil. The phase detection circuit may measure the current phase from a change in voltage applied to the resistor. 
     The wireless power feeder may further include: a first waveform rectifier that converts an analog waveform having the same phase as that of a current waveform of the AC power into a digital waveform; and a second waveform rectifier that converts an analog waveform having the same phase as that of a voltage waveform of the AC power into a digital waveform. The phase detection circuit compares the edges of two digital waveforms to detect the phase difference. The digitization makes clear the reference point used for comparing a current waveform and a voltage waveform, making it easy for the phase detection circuit to identify the phase difference. 
     A wireless power feeder according to a fourth aspect of the present invention also feeds power by wireless from a feeding coil to a receiving coil at the resonance frequency of the feeding coil and receiving coil. The wireless power feeder includes: a power circuit; a feeding coil; an exciting coil that is magnetically coupled to the feeding coil and feeds AC power fed from the power circuit to the feeding coil; a detection coil that generates inductive current using a magnetic field generated by the AC power; and a phase detection circuit that detects the phase difference between the voltage phase and current phase of the AC power. The power circuit includes first and second current paths and makes first and second switches connected in series respectively to the first and second current paths alternately conductive to feed the AC power to the exciting coil. The phase detection circuit measures the phase of the inductive current flowing in the detection coil to achieve measurement of the current phase of the AC power. 
     A wireless power feeder according to a fifth aspect of the present invention also feeds power by wireless from a feeding coil to a receiving coil at the resonance frequency of the feeding coil and receiving coil. The wireless power feeder includes: a power circuit that feeds AC power to the feeding coil at the drive frequency; a feeding coil circuit that includes the feeding coil and a capacitor and resonate at the resonance frequency; a detection coil that generates inductive current using a magnetic field generated by the AC power of the feeding coil circuit; and a phase detection circuit that detects the phase difference between the voltage phase and current phase of the AC power. The power circuit includes first and second current paths and makes first and second switches connected in series respectively to the first and second current paths alternately conductive to feed the AC power to the feeding coil circuit. The phase detection circuit measures the phase of inductive current passing through the detection coil to achieve measurement of the current phase of the AC power. 
     In such a configuration, when the drive frequency of the power circuit and resonance frequency are made coincide with each other, the power transmission efficiency in the entire system can be enhanced. The current phase is measured from the inductive current of the detection coil, so that a measurement load is not directly applied to the feeding coil. 
     The detection coil may generate the inductive current using a magnetic field generated by the AC power flowing through the feeding coil or may generate the inductive current using a magnetic field generated by the AC power flowing through the exciting coil. 
     The wireless power feeder also may further include a drive frequency tracking circuit that adjusts the drive frequency so as to reduce the detected phase difference to allow the drive frequency to track the resonance frequency. In this case, it is possible to allow the drive frequency to track the resonance frequency, thereby making it easy to maintain high power transmission efficiency. 
     The detection coil may be wounded around a toroidal core. A part of the feeding coil or exciting coil may be made to pass through the toroidal core to constitute a coupling transformer by one of the feeding and exciting coils and detection coil. Further, the phase detection circuit may measure the current phase from a change in voltage applied to a resistor connected in parallel to both ends of the detection coil. 
     This wireless power feeder also may further include: a first waveform rectifier that converts an analog waveform having the same phase as that of a current waveform of the AC power into a digital waveform; and a second waveform rectifier that converts an analog waveform having the same phase as that of a voltage waveform of the AC power into a digital waveform. The phase detection circuit may compare the edges of two digital waveforms to detect the phase difference. 
     A wireless power transmission system according to the present invention includes: one of the wireless power feeders described above; a receiving coil; and a loading coil that is magnetically coupled to the receiving coil and receives power that the receiving coil has received from the feeding coil. 
     It is to be noted that any arbitrary combination of the above-described structural components and expressions changed between a method, an apparatus, a system, etc. are all effective as and encompassed by the present embodiments. 
     According to the present invention, it is possible to detect the phase of supply power while suppressing influence on the resonance characteristics in a wireless power feeding technique of a magnetic resonance type. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above features and advantages of the present invention will be more apparent from the following description of certain preferred embodiments taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a basic system configuration view of a wireless power transmission system; 
         FIG. 2  is a view illustrating a current path formed when a first switching transistor is turned conductive; 
         FIG. 3  is a view illustrating a current path formed when a second switching transistor is turned conductive; 
         FIG. 4  is a time chart illustrating the voltage/current changing process in two switching transistors at the resonance time; 
         FIG. 5  is a graph illustrating a relationship between the impedance of a feeding coil circuit and drive frequency; 
         FIG. 6  is a graph illustrating a relationship between the output power efficiency and drive frequency; 
         FIG. 7  is a time chart illustrating the voltage/current changing process in the switching transistor observed in the case where the drive frequency is higher than resonance frequency; 
         FIG. 8  is a time chart illustrating the voltage/current changing process in the switching transistor observed in the case where the drive frequency is lower than resonance frequency; 
         FIG. 9  is a system configuration view of a wireless power transmission system according to the first embodiment; 
         FIG. 10  is a time chart illustrating the changing process of various voltages input to the phase detection circuit; 
         FIG. 11  is a graph illustrating a relationship between control voltage and drive frequency; 
         FIG. 12  is a system configuration view of the wireless power transmission system which is a modification of the first embodiment; 
         FIG. 13  is a system configuration view of a wireless power transmission system according to a second embodiment; 
         FIG. 14  is an enlarged configuration view of a detection coil and a feeding coil; 
         FIG. 15  is an equivalent circuit diagram of a coupling transformer constituted by a detection coil and a feeding coil; 
         FIG. 16  is a graph illustrating a relationship between the impedance Z of a resonance circuit and drive frequency; 
         FIG. 17  is a graph illustrating a relationship between the output power efficiency and drive frequency; 
         FIG. 18  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency and resonance frequency coincide with each other; 
         FIG. 19  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency is higher than the resonance frequency; 
         FIG. 20  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency is lower than the resonance frequency; 
         FIG. 21  is a time chart illustrating the changing process of various voltages input to the phase detection circuit; 
         FIG. 22  is a graph illustrating a relationship between control voltage and drive frequency; 
         FIG. 23  is a system configuration view of the wireless power transmission system according to a first modification of the second embodiment; 
         FIG. 24  is a system configuration view of the wireless power transmission system according to a second modification of the second embodiment; 
         FIG. 25  is a system configuration view of the wireless power transmission system according to a third modification of the second embodiment; 
         FIG. 26  is a system configuration view of a wireless power transmission system according to a third embodiment; 
         FIG. 27  is a system configuration view of a wireless power transmission system according to a fourth embodiment; 
         FIG. 28  is a system configuration view of the wireless power transmission system according to a first modification of the fourth embodiment; and 
         FIG. 29  is a system configuration view of the wireless power transmission system according to a second modification of the fourth embodiment. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     Preferred embodiments of the present invention will be explained below in detail with reference to the accompanying drawings. A wireless transmission system  300  in the present embodiment has not only a wireless power feeding function but also an automatic drive frequency tracking function. 
     First Embodiment 
     Push-Pull Type 
       FIG. 1  is a system configuration view of a wireless power transmission system  100  without automatic drive frequency tracking function. The wireless power transmission system  100  includes a power circuit  200 , an exciting circuit  110 , a feeding coil circuit  120 , a receiving coil circuit  130 , and a loading circuit  140 . A distance of several meters is provided between the feeding coil circuit  120  and receiving coil circuit  130 . The wireless power transmission system  100  mainly aims to feed power from the feeding coil circuit  120  to receiving coil circuit  130  by wireless. 
     The wireless power transmission system  100  illustrated in  FIG. 1  is assumed to operate at ISM (Industry-Science-Medical) frequency band. The following description will be made assuming that the resonance frequency fr of the feeding coil circuit  120  or receiving coil circuit  130  is 13.56 MHz within the ISM frequency band. 
     The exciting circuit  110  is a circuit in which an exciting coil L 1  and a transformer T 2  secondary coil Li are connected in series. The exciting circuit  110  receives AC power from the power circuit  200  through the transformer T 2  secondary coil Li. The transformer T 2  secondary coil Li constitutes a coupling transformer T 2  together with a transformer T 2  primary coil Ld and a transformer T 2  primary coil Lb and receives AC power by electromagnetic induction. The number of windings of the exciting coil L 1  is 1, diameter of the wire of the exciting coil L 1  is 3 mm, and diameter of the exciting coil L 1  itself is 210 mm. Current I 1  flowing in the exciting circuit  110  is AC. The direction of an arrow in the diagram of the exciting circuit  110  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. 
     The feeding coil circuit  120  is a circuit in which a feeding coil L 2  and a capacitor C 2  are connected in series. The exciting coil L 1  and feeding coil L 2  face each other. The distance between the exciting coil L 1  and feeding coil L 2  is as comparatively small as 10 mm or less. Thus, the exciting coil L 1  and feeding coil L 2  are electromagnetically strongly coupled to each other. The number of windings of the feeding coil L 2  is 7, diameter of the wire of the feeding coil L 2  is 5 mm, and diameter of the feeding coil L 2  itself is 280 mm. When the current I 1  is made to flow in the exciting coil L 1 , an electromotive force occurs in the feeding coil circuit  120  to cause current i 2  to flow in the feeding coil circuit  120 . The direction of an arrow in the diagram of the feeding coil circuit  120  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current I 1  and current I 2  are opposite (having opposite phases). The magnitude of the current I 2  is significantly larger than that of the current I 1 . The values of the feeding coil L 2  and capacitor C 2  are set such that the resonance frequency fr of the feeding coil circuit  120  is 13.56 MHz. 
     The receiving coil circuit  130  is a circuit in which a receiving coil L 3  and a capacitor C 3  are connected in series. The feeding coil L 2  and receiving coil L 3  face each other. The distance between the feeding coil L 2  and receiving coil L 3  is as comparatively large as about 0.2 m to 1 m. The number of windings of the receiving coil L 3  is 7, diameter of the wire of the receiving coil L 3  is 5 mm, and diameter of the receiving coil L 3  itself is 280 mm. The values of the receiving coil L 3  and capacitor C 3  are set such that the resonance frequency fr of the receiving coil circuit  130  is also 13.56 MHz. The feeding coil L 2  and receiving coil L 3  need not have the same shape. When the feeding coil circuit  120  generates a magnetic field at the resonance frequency fr, the feeding coil circuit  120  and receiving coil circuit  130  magnetically resonate, causing large current I 3  to flow in the receiving coil circuit  130 . The direction of an arrow in the diagram of the receiving coil circuit  130  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current I 2  and current I 3  are opposite (having opposite phases). That is, the current I 1  and current I 3  are in-phase. 
     The loading circuit  140  is a circuit in which a loading coil L 4  and a load R are connected in series. The receiving coil L 3  and loading coil L 4  face each other. The distance between the receiving coil L 3  and loading coil L 4  is as comparatively small as about 10 mm or less. Thus, the receiving coil L 3  and loading coil L 4  are electromagnetically strongly coupled to each other. In the present embodiment, the number of windings of the loading coil L 4  is 1, diameter of the wire of the loading coil L 4  is 3 mm, and diameter of the loading coil L 4  itself is 210 mm. When the current I 3  is made to flow in the receiving coil L 3 , an electromotive force occurs in the loading circuit  140  to cause current I 4  to flow in the loading circuit  140 . The direction of an arrow in the diagram of the loading circuit  140  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current I 3  and current I 4  are opposite (having opposite phases). That is, the current I 2  and current I 4  are in-phase. 
     The AC power fed from the power circuit  200  is transmitted by the exciting circuit  110  and feeding coil circuit  120 , received by the receiving coil circuit  130  and loading circuit  140 , and taken from the load R. The frequencies of the currents I 1  to I 4  flowing in the exciting circuit  110 , feeding coil circuit  120 , receiving coil circuit  130 , and loading circuit  140  are the same. 
     If the load R is connected in series to the receiving coil circuit  130 , the Q-value of the receiving coil circuit  130  is degraded. Therefore, the receiving coil circuit  130  for power reception and loading circuit  140  for power extraction are separated from each other. In order to enhance the power transmission efficiency, the center lines of the exciting coil L 1 , feeding coil L 2 , receiving coil L 3 , and loading coil L 4  are preferably made to coincide with one another. 
     The power circuit  200  is a push-pull circuit operating at a drive frequency fo and has a vertically symmetrical configuration as illustrated in  FIG. 1 . The exciting circuit  110  receives AC power at the drive frequency fo from the power circuit  200 . In this case, the currents I 1  to I 4  at the drive frequency fo flow in the exciting circuit  110 , feeding coil circuit  120 , receiving coil circuit  130 , and loading circuit  140 . When the drive frequency fo and resonance frequency fr coincide with each other, that is, when the drive frequency fo assumes 13.56 MHz, the feeding coil circuit  120  and receiving coil circuit  130  magnetically resonate, maximizing the power transmission efficiency. 
     An oscillator  202  is connected to the primary side of a gate-drive transformer T 1  included in the power circuit  200 . The oscillator  202  generates AC voltage at the drive frequency fo. Although the voltage waveform may be a sine wave, it is assumed here that the voltage waveform is a rectangular wave. The AC voltage causes current to flow in a transformer T 1  primary coil Lh alternately in both positive and negative directions. The transformer T 1  primary coil Lh, transformer T 1  secondary coil Lg, and transformer T 1  secondary coil Lf constitute a gate-drive coupling transformer T 1 . Electromagnetic induction causes current to flow also in the transformer T 1  secondary coil Lg and transformer T 1  secondary coil Lf alternately in both positive and negative directions. 
     The secondary coil of the transformer T 1  is center-point grounded. That is, one ends of the transformer T 1  secondary coil Lf and transformer T 1  secondary coil Lg are connected to each other and directly grounded. The other end of the transformer T 1  secondary coil Lf is connected to the gate of a switching transistor Q 1 , and the other end of the transformer T 1  secondary coil Lg is connected to the gate of a switching transistor Q 2 . The source of the switching transistor Q 1  and source of the switching transistor Q 2  are also grounded. Thus, when the oscillator  202  generates AC voltage of the drive frequency fo, voltage Vx (Vx&gt;0) of the drive frequency fo is applied alternately to the gates of the switching transistors Q 1  and Q 2 . As a result, the switching transistors Q 1  and Q 2  are alternately turned on/off at the drive frequency fo. 
     The switching transistors Q 1  and Q 2  are enhancement type MOSFET (Metal Oxide Semiconductor Field effect transistor) having the same characteristics but may be other transistors such as a bipolar transistor. Further, in the case where the drive frequency fo is lowered, other switches such as a relay switch may be used in place of the transistor. 
     Voltage between the source and drain of the switching transistor Q 1  is referred to as source-drain voltage VDS 1 , and voltage between the source and drain of the switching transistor Q 2  is referred to as source-drain voltage VDS 2 . Current flowing between the source and drain of the switching transistor Q 1  is referred to as source-drain current IDS 1 , and current flowing between the source and drain of the switching transistor Q 2  is referred to as source-drain current IDS 2 . The directions of arrows in the diagram of the power circuit  200  indicate the positive directions, and directions opposite to the directions of the arrows indicate the negative directions. 
     The drain of the switching transistor Q 1  is connected in series to a transformer T 2  primary coil Ld through an inductor Le and a capacitor Cb. Similarly, the drain of the switching transistor Q 2  is connected in series to a transformer T 2  primary coil Lb through an inductor Lc and a capacitor Ca. A smoothing inductor La and a power supply Vdd are connected to the connection point between the transformer T 2  primary coil Ld and transformer T 2  primary coil Lc. Further, a capacitor CQ 1  is connected in parallel to the source-drain of the switching transistor Q 1 , and a capacitor CQ 2  is connected in parallel to the source-drain of the switching transistor Q 2 . The inductors Le and Lc are coils having the same characteristics. The capacitors Cb and Ca are capacitors having the same characteristics, and capacitors CQ 1  and CQ 2  are capacitors having the same characteristics. 
     The inductor Le and capacitor Cb are inserted so as to shape the current waveform of the source-drain current IDS 1 , and inductor Lc and capacitor Ca are inserted so as to shape the current waveform of the source-drain current IDS 2 . Further, the capacitor CQ 1  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 1 , and capacitor CQ 2  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 2 . Even if the inductors Lc and Le, capacitors Ca, Cb, CQ 1 , and CQ 2  are omitted, the wireless power feeding using the power circuit  200  can be achieved. In particular, in the case where the drive frequency fo is low, it is easily possible to maintain the power transmission efficiency even if the inductors and capacitors are omitted. 
     The input impedance of the exciting circuit  110  is 50 (Ω). The number of windings of the transformer T 2  primary coil Lb and the number of windings of the transformer T 2  primary coil Ld are set such that the output impedance of the power circuit  200  is equal to the input impedance of 50(Ω). When the output impedance of the power circuit  200  and input impedance of the exciting circuit  110  coincide with each other, the power circuit  200  has the maximum output. 
       FIG. 2  is a view illustrating a current path formed when the switching transistor Q 1  is turned conductive. When the switching transistor Q 1  is turned conductive (ON), the switching transistor Q 2  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “first current path”) at this time is from the power supply Vdd through the smoothing inductor La, transformer T 2  primary coil Ld, capacitor Cb, inductor Le, and switching transistor Q 1  to the ground. The switching transistor Q 1  functions as a switch for controlling conduction/non-conduction of the first current path. 
       FIG. 3  is a view illustrating a current path formed when the switching transistor Q 2  is turned conductive. When the switching transistor Q 2  is turned conductive (ON), the switching transistor Q 1  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “second current path”) at this time is from the power supply Vdd through the smoothing inductor La, transformer T 2  primary coil Lb, capacitor Ca, inductor Lc, and switching transistor Q 2  to the ground. The switching transistor Q 2  functions as a switch for controlling conduction/non-conduction of the second current path. 
       FIG. 4  is a time chart illustrating the voltage/current changing process in the switching transistors Q 1  and Q 2 . Time period from time t 0  to time t 1  (hereinafter, referred to as “first time period”) is a time period during which the switching transistor Q 1  is ON while the switching transistor Q 2  is OFF. Time period from time t 1  to time t 2  (hereinafter, referred to as “second time period”) is a time period during which the switching transistor Q 1  is OFF while the switching transistor Q 2  is ON. Time period from time t 2  to time t 3  (hereinafter, referred to as “third time period”) is a time period during which the switching transistor Q 1  is ON while the switching transistor Q 2  is OFF. Time period from time t 3  to time t 4  (hereinafter, referred to as “fourth time period”) is a time period during which the switching transistor Q 1  is OFF while the switching transistor Q 2  is ON.  FIG. 4  illustrates waveforms observed in the case where the drive frequency fo and resonance frequency fr coincide with each other and the feeding coil circuit  120  and receiving coil circuit  130  are in a resonance state. 
     When the gate-source voltage VGS 1  of the switching transistor Q 1  exceeds a predetermined threshold, the switching transistor Q 1  is in a saturated state. Thus, when the switching transistor Q 1  is turned ON (conductive) at time t 0  which is the start timing of the first time period, the source-drain current IDS 1  starts flowing in the first current path illustrated in  FIG. 2 . Since current resonance occurs in the inductor Le and capacitor Cb inserted into the first current path, the current waveform of the source-drain current IDS 1  in the first time period does not assume a rectangular wave but the rising and falling edges become slower. 
     When the switching transistor Q 1  is turned OFF (non-conductive) at time t 1  which is the start timing of the second time period, the source-drain current IDS 1  does not flow. Since the capacitor CQ 1  is connected in parallel between the source and drain of the switching transistor Q 1 , the voltage waveform of the source-drain voltage VDS 1  in the second time period does not assume a rectangular wave but the rising and falling edges become slower. 
     Since the switching transistor Q 2  is OFF in the first time period, changes of the VGS 2 , IDS 2 , and VDS 2  in the first time period are the same as those of the VGS 1 , IDS 1 , and VDS 1  in the second time period. Since the switching transistor Q 2  is ON in the second time period, changes of the VGS 2 , IDS 2 , and VDS 2  in the second time period are the same as those of the VGS 1 , IDS 1 , and VDS 1  in the first period. In the third, fourth, and subsequent time periods, the same waveforms as in the first and second time periods are repeated. 
       FIG. 5  is a graph illustrating a relationship between the impedance Z of the feeding coil circuit  120  and drive frequency fo. The vertical axis represents the impedance Z of the feeding coil circuit  120 . The horizontal axis represents the drive frequency fo. The feeding coil circuit  120  is an LC circuit, so that the impedance Z of the feeding coil circuit  120  in view of the power circuit  200  or exciting circuit  110  is a minimum value Zmin at the resonance state. Although Zmin=0 at the resonance state is ideal, Zmin does not become zero in general since some resistance components are included in the feeding coil circuit  120 . 
     In  FIG. 5 , when the drive frequency fo is 13.56 MHz, that is, when the drive frequency fo and resonance frequency fr coincide with each other, the impedance Z becomes minimum and the feeding coil circuit  120  is in a resonance state. In a resonance state, the capacitive reactance and inductive reactance of the feeding coil circuit  120  cancel each other. When the drive frequency fo and resonance frequency fr deviate from each other, one of the capacitive reactance and inductive reactance prevails the other, so that the impedance Z is also increased. 
     To summarize, when the drive frequency fo of the power circuit  200  coincides with the resonance frequency fr, the AC current I 1  flows in the exciting circuit  110  at the resonance frequency fr. As a result, the current I 2  flows in the feeding coil circuit  120  at the resonance frequency fr, and the current I 3  flows in the receiving coil circuit  130  at the resonance frequency fr. The feeding coil L 2  and capacitor C 2  of the feeding coil circuit  120  and the receiving coil L 3  and capacitor C 3  of the receiving coil circuit  130  resonate at the same resonance frequency fr, so that the power transmission efficiency from the feeding coil L 2  to receiving coil L 3  becomes maximum. 
     When the drive frequency fo of the power circuit  200  and resonance frequency fr deviate from each other, the AC current I 1  flows in the exciting circuit  110  at a non-resonance frequency. Thus, the feeding coil circuit  120  or receiving coil circuit  130  is not in a resonance state, the power transmission efficiency is rapidly degraded. 
       FIG. 6  is a graph illustrating a relationship between the output power efficiency and drive frequency fo. The output power efficiency is a ratio of power actually fed from the feeding coil circuit  120  relative to the maximum output value. When the drive frequency fo coincides with the resonance frequency fr, a difference between the current phase and voltage phase becomes zero and therefore the power transmission efficiency becomes maximum, with the result that output power efficiency of 100(%) can be obtained. The output power efficiency can be measured from the magnitude of power taken from the load R. 
     As can be seen from the graph of  FIG. 6 , when the drive frequency fo is set to 14.06 MHz under the condition that the resonance frequency fr is 13.56 MHz, the output power efficiency is reduced to about 65(%). That is, the drive frequency fo and resonance frequency fr deviate from each other by 0.5 MHz, the power transmission efficiency is reduced by 35(%). 
       FIG. 7  is a time chart illustrating the voltage/current changing process in the switching transistor Q 2  observed in the case where the drive frequency fo is higher than the resonance frequency fr. In the case where the drive frequency fo is higher than the resonance frequency fr, an inductive reactance component appears in the impedance Z of the feeding coil circuit  120 , and the current phase of the current I 2  of the feeding coil circuit  120  delays with respect to the voltage phase. As described above, the current I 2  of the feeding coil circuit  120  and current I 1  of the exciting circuit  110  have just opposite phases. Further, the current I 1  of the exciting circuit  110  and source-drain current IDS 2  of the switching transistor Q 2  flowing in the second current path have just opposite phases. Thus, by measuring the current waveform of the source-drain current IDS 2  passing through the switching transistor Q 2 , the phase of the current I 2  of the feeding coil circuit  120  can be detected. Then, by comparing the current waveform of the source-drain current IDS 2  and voltage waveform of the source-drain voltage VDS 2 , a phase difference td between the current phase and voltage phase in the supply power can be detected. 
     As illustrated in  FIG. 4 , when the drive frequency fo coincides with the resonance frequency fr, the source-drain current IDS 2  starts flowing at time t 1  which is the start timing of the second time period. In this case, the phase difference td is 0. When the drive frequency fo is higher than the resonance frequency fr, the source drain current IDS 2  starts flowing at time t 5  which is later than time t 1 , so that the phase difference td (=t 1 −t 5 ) becomes less than 0. When the drive frequency fo and resonance frequency fr deviate from each other, the output power efficiency is degraded, and the source-drain current IDS 2  itself becomes reduced. 
       FIG. 8  is a time chart illustrating the voltage/current changing process in the switching transistor Q 2  observed in the case where the drive frequency fo is lower than the resonance frequency fr. In the case where the drive frequency fo is lower than the resonance frequency fr, a capacitive reactance component appears in the impedance Z, and the current phase of the current I 2  of the feeding coil circuit  120  advances with respect to the voltage phase. Thus, the source drain current IDS 2  starts flowing at time t 6  which is earlier than time t 1 . In this case, the phase difference td (=t 1 −t 6 ) is more than 0. The amplitude itself of the source-drain current IDS 2  becomes smaller than that at the resonance time. 
     The magnitude of the phase difference td and that of the deviation between the drive frequency fo and resonance frequency fr are proportional. Thus, by detecting the phase difference td and appropriately adjusting the drive frequency fo so as to eliminate the deviation between the drive frequency fo and resonance frequency fr, the resonance state can be maintained even if the resonance frequency fr is changed. 
       FIG. 9  is a system configuration view of a wireless power transmission system  300  according to the first embodiment. The wireless power transmission system  300  of the first embodiment has an “automatic drive frequency fo tracking function” in addition to the “wireless power feeding function” of the wireless power transmission system  100 . Components designated by the same reference numerals as those of  FIG. 1  have the same or corresponding functions as those in  FIG. 1 . In addition to the components illustrated in  FIG. 1 , the wireless power transmission system  300  further includes a first waveform rectifier  142 , a second waveform rectifier  144 , a phase detection circuit  150 , and a drive frequency tracking circuit  152 . Further, resistors R 1  to R 6  are added to a part of the power circuit  200 . 
     The wireless power transmission system  100  of  FIG. 1  as a basic configuration is assumed to operate at a predetermined resonance frequency fr. Thus, the drive frequency fr of the power circuit  200  is uniquely determined by the resonance frequency fr defined by the design of the feeding coil circuit  120  and receiving coil circuit  130 . 
     However, the resonance frequency fr slightly changes depending on use condition or use environment of the feeding coil circuit  120  or receiving coil circuit  130 . Further, in the case where the feeding coil circuit  120  or receiving coil circuit  130  is replaced with new one, the resonance frequency fr changes. Alternatively, there may be case where the resonance frequency needs to be changed aggressively by setting the electrostatic capacitance of the capacitor C 2  or capacitor C 3  variable. Even in such a case, the wireless power transmission system  300  can make the drive frequency fo and resonance frequency fr to automatically coincide with each other. 
     In the wireless power transmission system  300 , the resistor R 2  is connected in series between the source of the switching transistor Q 1  and ground, and resistor R 1  is connected in series between the source of the switching transistor Q 2  and ground. These resistors are referred to as “first resistors”. Further, the resistors R 4  and R 6  are connected in parallel to the source-drain of the switching transistor Q 1 , and resistors R 3  and R 5  are connected in parallel to the source-drain of the switching transistor Q 2 . A combination of the resistors R 4  and R 6  or combination of the resistors R 3  and R 5  is referred to as “second resistor”. The resistors R 1 , R 3 , and R 5  are equal respectively to the resistors R 2 , R 4 , and R 6  in the resistance value. 
     The phase difference td is measured based on potential Vp 1  (intermediate potential of the second resistor) of a connection point A between the resistors R 3  and R 5  and potential Vq 1  (voltage value applied to the first resistors) of a connection point B between the source of the switching transistor Q 2  and resistor R 1 . As described with reference to  FIGS. 7 and 8 , the voltage phase can be measured from the analog waveform of the source-drain voltage VDS 2 . In the wireless power transmission system  300 , the source-drain voltage VDS 2  is divided by the resistors R 3  and R 5 , and the potential Vp 1  is taken as the intermediate potential of the source-drain voltage VDS 2 . Even in the case where the source-drain voltage VDS 2  is increased, the voltage VDS 2  can be reduced to a manageable level by the voltage division. In the case where the source-drain voltage VDS 2  can be handled without modification, the voltage division need not be performed. 
     The voltage phase can be measured from elements other than the source-drain voltage VDS 2 . For example, the source-gate voltage VGS 2  or voltages of the both ends of the transformer T 1  primary coil may be set as a measurement target. 
     The current phase can be measured from the analog waveform of the source-drain current IDS 2 . The potential Vq 1  of the connection point B has the same phase as that of the source-drain current IDS 2 , so that the current phase can be measured from the analog waveform of the potential Vq 1 . By comparing the analog waveforms of the potential Vp 1  and potential Vq 1 , the phase difference td between the voltage phase and current phase can be identified. 
     Although the current phase and voltage phase are measured from the switching transistor Q 2  side in the wireless power transmission system  300 , the same results can be obtained by performing the measurement from the switching transistor Q 1  side. Further, although the resistors R 2 , R 4 , and R 6  are connected on the switching transistor Q 1  which is not set as a measurement target in order to make the circuit configuration of the power circuit  200  vertically symmetrical, these resistors may be omitted. 
     The potential Vp 1  and potential Vq 1  are digitized by the first waveform rectifier  142  and second waveform rectifier  144 , respectively. Although details will be described later with reference to  FIG. 10 , the first waveform rectifier  142  is an amplifier that outputs a saturated voltage Vp 2 =5 (V) when the potential Vp 1  exceeds a predetermined threshold, e.g., 0.1 (V). Thus, the potential Vp 1  of an analog waveform is converted to the voltage Vp 2  of a digital waveform by the first waveform rectifier  142 . Similarly, the second waveform rectifier  144  is an amplifier that outputs a saturated voltage Vq 2 =5 (V) when the voltage Vp 2  exceeds a predetermined threshold. The potential Vq 1  of an analog waveform is converted to the voltage Vq 2  of a digital waveform by the second waveform rectifier  144 . 
     The phase detection circuit  150  compares the digital waveforms of the voltage Vp 2  and voltage Vq 2  to calculate the phase difference td. The phase detection circuit  150  changes a control voltage Vt in accordance with the phase difference td. The drive frequency tracking circuit  152  adjusts the drive frequency fo of the oscillator  202  in accordance with the control voltage Vt. 
     The drive frequency tracking circuit  152  and oscillator  202  may be integrated as a VCO (Voltage Controlled Oscillator). Further, an amplifier may be provided at the rear stage of the VCO so as to amplify the voltage to be fed to the transformer T 1  primary coil Lh. 
       FIG. 10  is a time chart illustrating the changing process of various voltages input to the phase detection circuit  150 . The source-drain voltage VDS 2  changes in synchronization with ON/OFF of the switching transistor Q 2 . By dividing the source-drain voltage VDS 2  by the resistors R 3  and R 5 , the potential Vp 1  is detected at the connection point A. The potential Vp 1  has the same phase as that of the source-drain voltage VDS 2  and has a waveform in which the amplitude (peak voltage) is reduced. In the first and third time periods during which the switching transistor Q 2  is OFF, the source-drain voltage VDS 2  is more than 0, that is, the potential Vp 1  is more than 0. The first waveform rectifier  142  amplifies the potential Vp 1  of an analog waveform to thereby generate the potential Vq 1  of a digital waveform. 
     The potential Vq 1  of the connection point B changes in synchronization with the source-drain current IDS 2 .  FIG. 10  illustrates a state where the drive frequency fo is higher than the resonance frequency fr and the current phase delays with respect to the voltage phase. The second waveform rectifier  144  amplifies the potential Vq 1  of an analog waveform to thereby generate the voltage Vq 2  of a digital waveform. 
     The phase detection circuit  150  compares falling edge time t 1  of the voltage Vp 2  and rising edge time t 5  of the voltage Vq 2  and calculates the phase difference td by subtracting t 5  from t 1 . The conversion of the analog waveforms of the potentials Vp 1  and Vq 1  into digital waveforms using the first waveform rectifier  142  and second waveform rectifier  144  makes it easier for the phase detection circuit  150  to detect the phase difference td. As a matter of course, the phase detection circuit  150  may detect the phase difference td by directly comparing the potential Vp 1  and potential Vq 1 . 
     If the current i 2  flowing in the feeding coil L 2  is set as a measurement target as in the Patent Document 2, a new load is applied to the feeding coil circuit  120  to change the impedance Z of the feeding coil circuit  120 , resulting in degradation of the Q-value. Connecting the phase detection circuit  150  to the current path of the resonating feeding coil L 2  is like measuring the vibration of a tuning fork while touching the tuning fork. In the wireless power transmission system  300  according to the first embodiment, the current phase is measured based on the potential Vq 1  in the power circuit  200 . The measurement load is not applied to the four resonance circuits (exciting circuit  110 , feeding coil circuit  120 , receiving coil circuit  130 , and loading circuit  140 ), so that it is possible to measure the current phase while suppressing the influence on the Q-value. 
       FIG. 11  is a graph illustrating a relationship between the control voltage Vt and drive frequency fo. The relationship of  FIG. 11  is set in the drive frequency tracking circuit  152 . The phase difference td is proportional to the variation of the resonance frequency fr. Thus, the phase detection circuit  150  determines the variation of the control voltage Vt in accordance with the phase difference td, and the drive frequency tracking circuit  152  determines the drive frequency fo in accordance with the control voltage Vt. 
     The resonance frequency fr is 13.56 MHz in the initial state and, accordingly, the drive frequency fo is set to 13.56 MHz. The control voltage Vt is initially set to 3 (V). Here, it is assumed that the resonance frequency fr is changed from 13.56 MHz to 12.56 MHz. Since the drive frequency fo (=13.56 MHz) is higher than the resonance frequency fo (=12.56 MHz) in this state, the phase difference td is smaller than 0. The phase difference td is proportional to the variation (−1.0 MHz) of the resonance frequency fr. The phase detection circuit  150  determines the variation of the control voltage Vt based on the phase difference td. In this example, the phase detection circuit  150  sets the variation of the control voltage Vt to −1 (V) and outputs new control voltage Vt=2 (V). The drive frequency tracking circuit  152  outputs the drive frequency fo=12.56 MHs corresponding to the control voltage Vt=2 (V) according to the relationship represented by the graph of  FIG. 11 . With the above processing, it is possible to allow the drive frequency fo to automatically track a change of the resonance frequency fr. 
     The phase detection circuit  150 , the drive frequency tracking circuit  152 , and oscillator  202  may be implemented as one chip. The processing of the phase detection circuit  150  or drive frequency tracking circuit  152  may be performed by software. For example, setting information in which the phase difference td and variation of the drive frequency fo have been previously associated may be retained. In this case, the drive frequency fo is adjusted in accordance with the magnitude of the detected phase difference td. 
       FIG. 12  is a system configuration view of a wireless power transmission system  400  which is a modification of the wireless power transmission system of the first embodiment. In the wireless power transmission system  400 , the power circuit  200  directly drives the feeding coil circuit  120  without intervention of the exciting circuit  110 . Components designated by the same reference numerals as those of  FIG. 1  or  FIG. 9  have the same or corresponding functions as those in  FIG. 1  or  FIG. 9 . 
     The feeding coil circuit  120  of the wireless power transmission system  400  is a circuit in which the transformer T 2  secondary coil Li is connected in series to the feeding coil L 2  and capacitor C 2 . The transformer T 2  secondary coil Li constitutes the coupling transformer T 2  together with the transformer T 2  primary coil Lb and transformer T 2  primary coil Ld and receives AC power from the power circuit  200  by electromagnetic induction. As described above, the AC power may be directly fed from the power circuit  200  to the feeding coil circuit  200  without intervention of the exciting circuit  110 . 
     Second Embodiment 
     Half-Bridge Type 
       FIG. 13  is a system configuration view of a wireless power transmission system  1100  according to a second embodiment. The wireless power transmission system  1100  includes, as basic components, a power circuit  1200 , a receiving coil circuit  1130 , and a loading circuit  1140 . Further, the wireless power transmission system  1100  includes, as components for automatically adjusting the drive frequency fo, a first waveform rectifier  1142 , a second waveform rectifier  1144 , a phase detection circuit  1150 , and a drive frequency tracking circuit  1152 . The power circuit  1200  further includes a feeding coil L 2 . A distance of several meters is provided between the feeding coil L 2  and receiving coil circuit  1130 . The wireless power transmission system  1100  mainly aims to feed power from the feeding coil L 2  to receiving coil circuit  1130  by wireless. The wireless power transmission system  1100  according to the second embodiment is assumed to operate at around 100 kHz. Thus, the resonance frequency fr of the feeding coils L 2  and L 3  is set to 100 MHz. The wireless power transmission system according to the present embodiment can be made to operate at a high-frequency band such as an ISM (Industry-Science-Medical) frequency band. 
     The power circuit  1200  is a half-bridge type circuit that directly feeds AC power to the feeding coil L 2  without intervention of the exciting coil. As illustrated in  FIG. 13 , the power circuit  1200  has a vertically symmetrical configuration. Current IS flowing in the feeding coil L 2  is AC. The direction of an arrow in the diagram of the feeding coil L 2  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. In the present embodiment, the number of windings of the feeding coil L 2  is 7, diameter of the wire of the feeding coil L 2  is 5 mm, and diameter of the feeding coil L 2  itself is 280 mm. 
     The receiving coil circuit  1130  is a circuit in which a receiving coil L 3  and a capacitor C 3  are connected in series. The feeding coil L 2  and receiving coil L 3  face each other. The distance between the feeding coil L 2  and receiving coil L 3  is as comparatively large as about 0.2 m to 1 m. The number of windings of the receiving coil L 3  is 7, diameter of the wire of the receiving coil L 3  is 5 mm, and diameter of the receiving coil L 3  itself is 280 mm. The values of the receiving coil L 3  and capacitor C 3  are set such that the resonance frequency fr of the receiving coil circuit  1130  is also 100 kHz. The feeding coil L 2  and receiving coil L 3  need not have the same shape. When the feeding coil circuit  1120  generates a magnetic field at the resonance frequency fr, the feeding coil circuit  1120  and receiving coil circuit  1130  magnetically resonate, causing large current I 3  to flow in the receiving coil circuit  1130 . The direction of an arrow in the diagram of the receiving coil circuit  1130  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current I 2  and current I 3  are opposite (having opposite phases). 
     The loading circuit  1140  is a circuit in which a loading coil L 4  and a load R are connected in series. The receiving coil L 3  and loading coil L 4  face each other. The distance between the receiving coil L 3  and loading coil L 4  is as comparatively small as about 10 mm or less. Thus, the receiving coil L 3  and loading coil L 4  are electromagnetically strongly coupled to each other. In the present embodiment, the number of windings of the loading coil L 4  is 1, diameter of the wire of the loading coil L 4  is 3 mm, and diameter of the loading coil L 4  itself is 210 mm. When the current I 3  is made to flow in the receiving coil L 3 , an electromotive force occurs in the loading circuit  1140  to cause current I 4  to flow in the loading circuit  1140 . The direction of an arrow in the diagram of the loading circuit  140  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current I 3  and current I 4  are opposite (having opposite phases). The AC power transmitted from the feeding coil L 2  of the power circuit  1200  is received by the receiving coil circuit  1130  and loading circuit  1140 , and taken from the load R. 
     If the load R is connected in series to the receiving coil circuit  1130 , the Q-value of the receiving coil circuit  1130  is degraded. Therefore, the receiving coil circuit  1130  for power reception and loading circuit  1140  for power extraction are separated from each other. In order to enhance the power transmission efficiency, the center lines of the exciting coil L 1 , feeding coil L 2 , receiving coil L 3 , and loading coil L 4  are preferably made to coincide with one another. 
     A configuration of the power circuit  1200  will be described. An oscillator  1202  is connected to the primary side of the gate-drive transformer T 1 . The oscillator  1202  functions as a “power transmission control circuit” that generates AC voltage at the drive frequency fo. Although the voltage waveform may be a sine wave, it is assumed here that the voltage waveform is a rectangular wave. The AC voltage causes current to flow in the transformer T 1  primary coil Lh alternately in both positive and negative directions. The transformer T 1  primary coil Lh, transformer T 1  secondary coil Lg, and transformer T 1  secondary coil Lf constitute a gate-drive coupling transformer T 1 . Electromagnetic induction causes current to flow also in the transformer T 1  secondary coil Lg and transformer T 1  secondary coil Lf alternately in both positive and negative directions. 
     One end of the transformer T 1  secondary coil Lf is connected to the gate of the switching transistor Q 1 , and the other end thereof is connected to the source of the switching transistor Q 1 . One end of the transformer T 1  secondary coil Lg is connected to the gate of the switching transistor Q 2 , and the other end thereof is connected to the source of the switching transistor Q 2 . When the oscillator  1202  generates AC voltage of the drive frequency fo, voltage Vx (Vx&gt;0) of the drive frequency fo is applied alternately to the gates of the switching transistors Q 1  and Q 2 . As a result, the switching transistors Q 1  and Q 2  are alternately turned on/off at the drive frequency fo. The switching transistors Q 1  and Q 2  are enhancement type MOSFET (Metal Oxide Semiconductor Field effect transistor) having the same characteristics but may be other transistors such as a bipolar transistor. Other switches such as a relay switch may be used in place of the transistor. 
     The drain of the switching transistor Q 1  is connected to the positive terminal of a power supply Vdd 1 . The negative terminal of the power supply Vdd 1  is connected to the source of the switching transistor Q 1  through the capacitor C 1  and feeding coil L 2 . The voltage at the negative terminal of the power supply Vdd 1  is the ground voltage. The source of the switching transistor Q 2  is connected to the negative terminal of a power supply Vdd 2 . The positive terminal of the power supply Vdd 2  is connected to the drain of the switching transistor Q 2  through the capacitor C 1  and feeding coil L 2 . The voltage at the positive terminal of the power supply Vdd 2  is the ground voltage. 
     Voltage between the source and drain of the switching transistor Q 1  is referred to as source-drain voltage VDS 1 , and voltage between the source and drain of the switching transistor Q 2  is referred to as source-drain voltage VDS 2 . Current flowing between the source and drain of the switching transistor Q 1  is referred to as source-drain current IDS 1 , and current flowing between the source and drain of the switching transistor Q 2  is referred to as source-drain current IDS 2 . The directions of arrows in the diagram of the power circuit  1200  indicate the positive directions, and directions opposite to the directions of the arrows indicate the negative directions. 
     The values of the capacitor C 1  and feeding coil L 2  are set so as to resonate at the resonance frequency fr. In other words, the capacitor C 1  and feeding coil L 2  constitute a “resonance circuit” of the resonance frequency fr. Further, the existence of the capacitor C 1  and feeding coil L 2  makes the current waveforms of the source-drain current IDS 1  and source-drain current IDS 2  to be sine waveforms. 
     The capacitor CQ 1  is connected parallel to the source-drain of the switching transistor Q 1 , and capacitor CQ 2  is connected in parallel to the source-drain of the switching transistor Q 2 . The capacitors CQ 1  and CQ 2  have the same characteristics. The capacitor CQ 1  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 1 , and capacitor CQ 2  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 2 . Even if capacitors CQ 1  and CQ 2  are omitted, the wireless power feeding using the power circuit  1200  can be achieved. In particular, in the case where the drive frequency fo is low, the influence of the capacitors is reduced. 
     When the switching transistor Q 1  is turned conductive (ON), the switching transistor Q 2  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “first current path  1102 ”) at this time starts from the power supply Vdd 1 , passes through the switching transistor Q 1 , the feeding coil L 2  and the capacitor C 1  and returns to Vdd 1 . The switching transistor Q 1  functions as a switch for controlling conduction/non-conduction of the first current path. 
     When the switching transistor Q 2  is turned conductive (ON), the switching transistor Q 1  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “second current path  1104 ”) at this time starts from the power supply Vdd 2 , passed through the capacitor C 1 , the feeding coil L 2 , and switching transistor Q 2  and return to Vdd 2 . The switching transistor Q 2  functions as a switch for controlling conduction/non-conduction of the second current path. 
     When the oscillator  1202  feeds the AC voltage at the resonance frequency fr, a first current path  1102  and a second current path  1104  are alternately switched at the resonance frequency fr. Since the AC current of the resonance frequency fr flows in the capacitor C 1  and feeding coil L 2 , the capacitor C 1  and feeding coil L 2  are in a resonance state. The receiving coil circuit  1130  is also a resonance circuit of the resonance frequency fr, so that the feeding coil L 2  and receiving coil L 3  magnetically resonate. At this time, the maximum transmission efficiency can be obtained. 
     The resonance frequency fr slightly changes depending on use condition or use environment of the feeding coil circuit  1120  or receiving coil circuit  1130 . Further, in the case where the feeding coil circuit  120  or receiving coil circuit  130  is replaced with new one, the resonance frequency fr changes. Alternatively, there may be case where the resonance frequency needs to be changed aggressively by setting the electrostatic capacitance of the capacitor C 2  or capacitor C 3  variable. Even in such a case, the wireless power transmission system  1100  can make the drive frequency fo and resonance frequency fr to automatically coincide with each other. 
     In order to make the drive frequency fo to track the resonance frequency fr, the following configuration is added. Resistors R 1  and R 2  are connected to both ends of the oscillator  1202 . A connection point A between the resistors R 1  and R 2  is connected to the phase detection circuit  1150  through the second waveform rectifier  1144 . The phase detection circuit  1150  measures the voltage phase of the AC power fed by the power circuit  1200  based on the potential Vp 1  of the connection point A according to the following method. 
     The AC voltage generated by the oscillator  1202  is divided by the resistors R 1  and R 2 , and the potential Vp 1  is taken as the intermediate potential of the AC voltage. Even in the case where the AC voltage generated by the oscillator  1202  is large, the AC voltage can be reduced to a manageable level by the voltage division. In the case where the AC voltage generated by the oscillator  1202  can be handled without modification, the voltage division need not be performed. The voltage phase may be measured from the source-drain voltages VDS 1  and VDS 2  or source-gate voltages VGS 1  and VGS 2 . 
     A detection coil LSS is provided near the feeding coil L 2 . The detection coil LSS is a coil wounded around a core  1154  (toroidal core) having a penetration hole NS times. A part of the feeding coil L 2  penetrates the core  1154 , so that the feeding coil L 2  and detection coil LSS constitute a coupling transformer. Inductive current ISS is made to flow in the detection coil LSS by an AC magnetic field generated by AC current IS. The current IS and inductive current ISS have the same phase. 
     A resistor R 3  is connected to both ends of the detection coil LSS. One end B of the resistor R 3  is grounded, and the other end C thereof is connected to the phase detection circuit  1150  through the first waveform rectifier  1142 . The phase detection circuit  1150  measures the current phase of the AC power fed by the power circuit  1200  based on the potential Vq 1  of the connection point C according to the following method. The current IS and inductive current ISS have the same phase, and the inductive current ISS and potential Vq 1  have the same phase. Therefore, the current phase of the current IS can be measured from the voltage phase of the potential Vq 1 . By comparing the voltage waveforms of the potential Vp 1  and potential Vq 1 , the deviation between the voltage phase and current phase can be detected. 
     The potential Vp 1  and potential Vq 1  are digitized by the first waveform rectifier  1142  and second waveform rectifier  1144 , respectively. Although details will be described later with reference to  FIG. 21 , the first waveform rectifier  1142  is an amplifier that outputs a saturated voltage Vp 2 =5 (V) when the potential Vp 1  exceeds a predetermined threshold, e.g., 0.1 (V). Thus, even in the case where the potential Vp 1  assumes an analog waveform, the potential Vp 1  is converted into voltage Vp 2  of a digital waveform by the first waveform rectifier  1142 . The first waveform rectifier  1142  functions particularly effectively when the oscillator  1202  generates the AC voltage not of a rectangular waveform but of an analog waveform such as a sine wave. The second waveform rectifier  1144  is an amplifier that outputs a saturated voltage Vq 2 =5 (V) when the voltage Vp 2  exceeds a predetermined threshold. The potential Vq 1  of an analog waveform is converted to the voltage Vq 2  of a digital waveform by the second waveform rectifier  1144 . 
     The phase detection circuit  1150  compares the digital waveforms of the voltage Vp 2  and voltage Vq 2  to calculate the phase difference td. The phase detection circuit  1150  changes a control voltage Vt in accordance with the phase difference td. The drive frequency tracking circuit  1152  adjusts the drive frequency fo of the oscillator  1202  in accordance with the control voltage Vt. 
     The drive frequency tracking circuit  1152  and oscillator  1202  may be integrated as a VCO (Voltage Controlled Oscillator). Further, an amplifier may be provided at the rear stage of the VCO so as to amplify the voltage to be fed to the transformer T 1  primary coil Lh. 
       FIG. 14  is an enlarged configuration view of the detection coil LSS and feeding coil L 2 .  FIG. 14  illustrates a configuration around the detection coil LSS in detail. The core  1154  has a cylindrical shape having a penetration hole and is formed of a known material such as ferrite, silicon steel, or permalloy. The number of windings NS of the detection coil LSS in the present embodiment is 100. A part of the feeding coil L 2  penetrates the penetration hole of the core  1154 . This means that the number of windings NP of the feeding coil L 2  with respect to the core  1154  is one. With the above configuration, the detection coil LSS and feeding coil L 2  constitute a coupling transformer. 
       FIG. 15  is an equivalent circuit diagram of the coupling transformer constituted by the detection coil LSS and feeding coil L 2 . The feeding coil L 2  is on the primary side, and the detection coil LSS is on the secondary side, whereby the coupling transformer is formed therebetween. An AC magnetic field generated by the AC current IS of the feeding coil L 2  causes inductive current ISS having the same phase as that of the current IS to flow in the detection coil LSS. The magnitude of the inductive current ISS is represented by IS·(NP/NS) according to the law of equal ampere-turn. The potential Vq 1  at one end C of the detection coil LSS is set as a measurement target. The other end B of the detection coil LSS is grounded, so that the potential Vq 1  is equal to the voltage value applied to the resistor R 3 . 
       FIG. 16  is a graph illustrating a relationship between the impedance Z of the resonance circuit and drive frequency fo. The vertical axis represents the impedance Z of the resonance circuit part (series circuit of the capacitor C 1  and feeding coil L 2 ) in the power circuit  1200 . The horizontal axis represents the drive frequency fo. The impedance Z of the resonance circuit is a minimum value Zmin at the resonance state. Although Zmin=0 at the resonance state is ideal, Zmin does not become zero in general since some resistance components are included in the resonance circuit. 
     In  FIG. 16 , when the drive frequency fo is 100 kHz, that is, when the drive frequency fo and resonance frequency fr coincide with each other, the impedance Z becomes minimum and the capacitor C 1  and the feeding coil L 2  are in a resonance state. When the drive frequency fo and resonance frequency fr deviate from each other, one of the capacitive reactance and inductive reactance prevails the other, so that the impedance Z is also increased. 
     When the drive frequency fo of the power circuit  1200  coincides with the resonance frequency fr, the AC current IS flows in the feeding coil L 2  at the resonance frequency fr, and current I 3  flows in the receiving coil circuit  1130  at the resonance frequency fr. The combination of feeding coil L 2  and capacitor C 2 , and the receiving coil L 3  and capacitor C 3  of the receiving coil circuit  130  resonate at the same resonance frequency fr, so that the power transmission efficiency from the feeding coil L 2  to receiving coil L 3  becomes maximum. 
     When the drive frequency fo and resonance frequency fr deviate from each other, the AC current IS flows in the feeding coil L 2  at a non-resonance frequency. Thus, the feeding coil L 2  and the receiving coil L 3  are not in a resonance state, the power transmission efficiency is rapidly degraded. 
       FIG. 17  is a graph illustrating a relationship between the output power efficiency and drive frequency fo. The output power efficiency is a ratio of power actually fed from the feeding coil L 2  relative to the maximum output value. When the drive frequency fo coincides with the resonance frequency fr, a difference between the current phase and voltage phase becomes zero and therefore the power transmission efficiency becomes maximum, with the result that output power efficiency of 100(%) can be obtained. The output power efficiency can be measured from the magnitude of power taken from the load R. 
     As can be seen from the graph of  FIG. 17 , when the drive frequency fo is set to 105 kHz under the condition that the resonance frequency fr is 100 kHz, the output power efficiency is reduced to about 75(%). That is, the drive frequency fo and resonance frequency fr deviate from each other by 5 kHz, the power transmission efficiency is reduced by 25(%). 
       FIG. 18  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency fo and resonance frequency fr coincide with each other. Time period from time t 0  to time t 1  (hereinafter, referred to as “first time period”) is a time period during which the switching transistor Q 1  is ON while the switching transistor Q 2  is OFF. Time period from time t 1  to time t 2  (hereinafter, referred to as “second time period”) is a time period during which the switching transistor Q 1  is OFF while the switching transistor Q 2  is ON. Time period from time t 2  to time t 3  (hereinafter, referred to as “third time period”) is a time period during which the switching transistor Q 1  is ON while the switching transistor Q 2  is OFF. Time period from time t 3  to time t 4  (hereinafter, referred to as “fourth time period”) is a time period during which the switching transistor Q 1  is OFF while the switching transistor Q 2  is ON. 
     When the gate-source voltage VGS 1  of the switching transistor Q 1  exceeds a predetermined threshold, the switching transistor Q 1  is in a saturated state. Thus, when the switching transistor Q 1  is turned ON (conductive) at time t 0  which is the start timing of the first time period, the source-drain current IDS 1  starts flowing. In other words, the current IS starts flowing in the positive direction (in the first current path  1102 ). Current resonance occurs in the resonance circuit (feeding coil L 2  and capacitor C 1 ), so that the current waveform of the current IS in the first time period does not assume a rectangular waveform but the rising and falling edges become slower. 
     When the switching transistor Q 1  is turned OFF (non-conductive) at time t 1  which is the start timing of the second time period, the source-drain current IDS 1  does not flow. Instead, the switching transistor Q 2  is turned ON (conductive), the source-drain current IDS 2  starts flowing. That is, the current IS starts flowing in the negative direction (second current path  1104 ). 
     The current IS and inductive current ISS have the same phase, and potential Vq 1  and inductive current ISS have the same phase. Therefore, the current waveform of the current IS and voltage waveform of the potential Vq 1  synchronizes with each other. By observing the voltage waveform of the potential Vq 1 , the current phase of the current IS (source-drain currents IDS 1  and IDS 2 ) can be measured. In the third, fourth, and subsequent time periods, the same waveforms as in the first and second time periods are repeated. 
       FIG. 19  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency fo is higher than the resonance frequency fr. In the case where the drive frequency fo is higher than the resonance frequency fr, an inductive reactance component appears in the impedance Z of the resonance circuit, and the current phase of the current IS delays with respect to the voltage phase. As described above, the current IS and potential Vq 1  have the same phase, so that by comparing the voltage waveforms of the potential Vp 1  and potential Vq 1 , the phase difference td between the current phase and voltage phase in the supply power can be detected. 
     As illustrated in  FIG. 18 , when the drive frequency fo coincides with the resonance frequency fr, the current IS starts flowing at time t 1  which is the start timing of the second time period, and the potential Vq 1  becomes more than 0. In this case, the phase difference td is 0. When the drive frequency fo is higher than the resonance frequency fr, the current IS starts flowing at time t 5  which is later than time t 1 , and Vq 1  becomes more than 0, so that the phase difference td (=t 1 −t 5 ) becomes less than 0. When the drive frequency fo and resonance frequency fr deviate from each other, the output power efficiency is degraded, and the amplitude of the current IS or potential Vq 1  becomes smaller than that at the resonance state. 
       FIG. 20  is a time chart illustrating the voltage/current changing process observed in the case where the drive frequency fo is lower than the resonance frequency fr. In the case where the drive frequency fo is lower than the resonance frequency fr, a capacitive reactance component appears in the impedance Z, and the current phase of the current IS advances with respect to the voltage phase. The current IS starts flowing at time t 6  which is earlier than time t 1 , so that the phase difference td (=t 1 −t 6 ) becomes more than 0. The amplitude of the current IS or potential Vq 1  becomes smaller than that at the resonance time. 
       FIG. 21  is a time chart illustrating the changing process of various voltages input to the phase detection circuit  1150 . The potential Vp 1  changes in synchronization with the AC voltage of the oscillator  1202 . The potential Vp 1  is more than 0 in the first and third time periods. The first waveform rectifier  1142  is an amplifier that outputs a saturated voltage of 5 (V) when the potential Vp 1  exceeds a predetermined threshold, e.g., 0.1 (V). Thus, even in the case where the potential Vp 1  assumes an analog waveform, the first waveform rectifier  1142  can generate the voltage Vp 2  of a digital waveform. 
     The potential Vq 1  changes in synchronization with the current IS.  FIG. 21  illustrates a waveform observed in the case where the drive frequency fo is lower than the resonance frequency fr. Thus, the current phase advances with respect to the voltage phase. The second waveform rectifier  1144  amplifies the potential Vq 1  of an analog waveform to thereby generate the voltage Vq 2  of a digital waveform. 
     The phase detection circuit  1150  compares rising edge time t 0  of the voltage Vp 2  and rising edge time t 6  of the voltage Vq 2  and calculates the phase difference td by subtracting t 6  from t 0 . The conversion of the analog waveforms of the potentials Vp 1  and Vq 1  into digital waveforms using the first waveform rectifier  1142  and second waveform rectifier  1144  makes it easier for the phase detection circuit  1150  to detect the phase difference td. As a matter of course, the phase detection circuit  1150  may detect the phase difference td by directly comparing the potential Vp 1  and potential Vq 1 . 
     If the current IS flowing in the feeding coil L 2  is set as a measurement target as in the Patent Document 2, a new load is applied to the feeding coil L 2  to change the impedance Z of the resonance circuit, resulting in degradation of the Q-value. Connecting the phase detection circuit  1150  to the current path of the resonating feeding coil L 2  is like measuring the vibration of a tuning fork while touching the tuning fork. In the wireless power transmission system  1100  according to the second embodiment, the AC magnetic field generated by the feeding coil L 2  is utilized to cause the detection coil LSS to generate the inductive current ISS, whereby the current phase is measured. The measurement load is not applied to the power circuit  1200 , in particular, the resonance circuit part of the power circuit  1200 , so that it is possible to measure the current phase while suppressing the influence on the Q-value. 
     It is possible to use not only the feeding coil L 2  but also the receiving coil L 3  or loading coil L 4  as the primary coil to constitute a coupling transformer so as to cause the detection coil LSS to generate the inductive current ISS. 
       FIG. 22  is a graph illustrating a relationship between the control voltage Vt and drive frequency fo. The relationship of  FIG. 22  is set in the drive frequency tracking circuit  1152 . The phase difference td is proportional to the variation of the resonance frequency fr. Thus, the phase detection circuit  1150  determines the variation of the control voltage Vt in accordance with the phase difference td, and the drive frequency tracking circuit  1152  determines the drive frequency fo in accordance with the control voltage Vt. 
     The resonance frequency fr is 100 kHz in the initial state and, accordingly, the drive frequency fo is set to 100 kHz. The control voltage Vt is initially set to 3 (V). Here, it is assumed that the resonance frequency fr is changed from 100 kHz to 90 kHz. Since the drive frequency fo (=100 kHz) is higher than the resonance frequency fo (=90 kHz) in this state, the phase difference td is smaller than 0. The phase difference td is proportional to the variation (−10 kHz) of the resonance frequency fr. The phase detection circuit  1150  determines the variation of the control voltage Vt based on the phase difference td. In this example, the phase detection circuit  1150  sets the variation of the control voltage Vt to −1 (V) and outputs new control voltage Vt=2 (V). The drive frequency tracking circuit  1152  outputs the drive frequency fo=100 kHs corresponding to the control voltage Vt=2 (V) according to the relationship represented by the graph of  FIG. 22 . With the above processing, it is possible to allow the drive frequency fo to automatically track a change of the resonance frequency fr. 
     The phase detection circuit  1150 , the drive frequency tracking circuit  1152 , and oscillator  1202  may be implemented as one chip. The processing of the phase detection circuit  1150  or drive frequency tracking circuit  1152  may be performed by software. For example, setting information in which the phase difference td and variation of the drive frequency fo have been previously associated may be retained. In this case, the drive frequency fo is adjusted in accordance with the magnitude of the detected phase difference td. 
       FIG. 23  is a system configuration view of the wireless power transmission system  1100  according to a first modification of the second embodiment. Components designated by the same reference numerals as those of  FIG. 13  have the same or corresponding functions as those in  FIG. 13 . The first modification includes a coupling transformer constituted by a primary coil Lj and a secondary coil Lk. That is, the resonance circuit constituted by the capacitor C 1  and feeding coil L 2  is physically separated from the power systems such as power supply Vdd 1 , power supply Vdd 2 , switching transistors Q 1  and Q 2 , etc. The AC power controlled by the oscillator  1202  is fed to the resonance circuit (capacitor C 1  and feeding coil L 2 ) through the coupling transformer. 
       FIG. 24  is a system configuration view of the wireless power transmission system  1100  according to a second modification of the second embodiment. Components designated by the same reference numerals as those of  FIG. 13  have the same or corresponding functions as those in  FIG. 13 . In the system configuration illustrated in  FIG. 13 , a coupling transformer is constituted by the feeding coil L 2  and detection coil LSS that share the core  1154 . In the system configuration illustrated in  FIG. 24 , the current phase is measured using a detection coil circuit  1170 . The detection coil circuit  1170  does not share the core  1154  or the like with the power circuit  1200 , thereby increasing installation flexibility. 
     The detection coil circuit  1170  is a circuit in which the detection coil LSS and resistor R 3  are connected in series. The detection coil circuit  1170  is installed such that a magnetic flux generated by the feeding coil L 2  passes through the detection coil LSS. As in the case of  FIG. 13 , one end B of the resistor R 3  is grounded, and the potential Vq 1  is detected from the other end C of the resistor R 3 . The AC magnetic field generated by the current IS flowing in the feeding coil L 2  causes the inductive current ISS to flow in the detection coil circuit  1170 . By measuring the potential Vq 1  generated by the inductive current ISS, the phase difference td between the voltage phase and current phase can be measured. 
     The purpose of installing the detection coil  1170  is not to receive power from the feeding coil L 2  but to measure the current phase of the AC power fed from the feeding coil L 2 . Thus, the size of the detection coil LSS can be made sufficiently smaller than that of the feeding coil L 2 . In the measurement of the phase difference td, the magnetic field by which the inductive current ISS is generated may be generated not only by the current IS flowing in the feeding coil L 2  but also by the current I 3  flowing in the receiving coil L 3  or current I 4  flowing in the loading coil L 4 . 
       FIG. 25  is a system configuration view of the wireless power transmission system  1100  according to a third modification of the second embodiment. Components designated by the same reference numerals as those of  FIGS. 13 ,  23 , and  24  have the same or corresponding functions as those in  FIGS. 13 ,  23 , and  24 . As in the case of the first modification, the third modification includes a coupling transformer constituted by the primary coil Lj and secondary coil Lk. The resonance circuit constituted by the capacitor C 1  and feeding coil L 2  is physically separated from the power systems such as power supply Vdd 1 , power supply Vdd 2 , switching transistors Q 1  and Q 2 , etc. The AC power controlled by the oscillator  1202  is fed to the resonance circuit (capacitor C 1  and feeding coil L 2 ) through the coupling transformer. 
     Third Embodiment 
     Half-Bridge Type 
       FIG. 26  is a system configuration view of a wireless power transmission system  1106  according to a third embodiment. In the wireless power transmission system  1100  according to the second embodiment, the oscillator  1202  directly drive the feeding coil L 2 ; while in the wireless power transmission system  1106  according to the third embodiment, the oscillator  1202  does not drive the feeding coil L 2  but drives the exciting coil L 1 . The other components of the wireless power transmission system  1106  are the same as those in  FIG. 13 , etc. Components designated by the same reference numerals as those of  FIG. 13 , etc. have the same or corresponding functions as those in  FIG. 13 , etc. 
     A power circuit  1204  feeds AC power to the exciting coil L 1  at the resonance frequency fr. The exciting coil L 1  and capacitor C 1  constitute a resonance circuit of the resonance frequency fr. A feeding coil circuit  1120  is a circuit in which the feeding coil L 2  and capacitor C 2  are connected in series. The exciting coil L 1  and feeding coil L 2  face each other. The distance between the exciting coil L 1  and feeding coil L 2  is as comparatively small as about 10 mm. Thus, the exciting coil L 1  and feeding coil L 2  are electromagnetically strongly coupled to each other. When current IS is made to flow in the exciting coil L 1 , an electromotive force occurs in the feeding coil circuit  1120  to cause current I 2  to flow in the feeding coil circuit  1120 . The direction of an arrow in the diagram of the feeding coil circuit  1120  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. The flowing directions of the current IS and current I 2  are opposite (having opposite phases). The magnitude of the current I 2  is significantly larger than that of the current IS. The values of the feeding coil L 2  and capacitor C 2  are set such that the resonance frequency fr of the feeding coil circuit  1120  is 100 kHz. 
     Also in the wireless power transmission system  1106  of the third embodiment, the resistors R 1  and R 2  are connected to both ends of the oscillator  1202 , and the voltage phase is measured from the potential Vp 1  at the connection point A between the resistors R 1  and R 2 . In the third embodiment, the detection coil LSS is provided near the exciting coil L 1 , and the detection coil LSS and exciting coil L 1  constitute a coupling transformer. Inductive current ISS is made to flow in the detection coil LSS by a magnetic field generated by AC current IS. The current phase is measured based on the inductive current ISS according to the same method as the first embodiment. 
     Also in the third embodiment, it is possible to use not only the exciting coil L 1  but also the feeding coil L 2 , receiving coil L 3  or loading coil L 4  as the primary coil to constitute a coupling transformer so as to cause the detection coil LSS to generate the inductive current ISS. The inductive current ISS may be generated using the detection coil circuit  1170  described with reference to  FIGS. 24 and 25 . 
     Fourth Embodiment 
     Push-Pull Type 
       FIG. 27  is a system configuration view of a wireless power transmission system  1108  according to a fourth embodiment. The wireless power transmission system  1108  includes a power circuit  1206 , an exciting circuit  1110 , a feeding coil circuit  1120 , a receiving coil circuit  1130 , and a loading circuit  1140 . A distance of several meters is provided between the feeding coil circuit  1120  and receiving coil circuit  1130 . The wireless power transmission system  1108  mainly aims to feed power from the feeding coil circuit  1120  to receiving coil circuit  1130  by wireless. Components designated by the same reference numerals as those of  FIG. 13  and  FIGS. 23 to 26  have the same or corresponding functions as those described above. 
     The exciting circuit  1110  is a circuit in which an exciting coil L 1  and a transformer T 2  secondary coil Li are connected in series. The exciting circuit  1110  receives AC power from the power circuit  1206  through the transformer T 2  secondary coil Li. The transformer T 2  secondary coil Li constitutes a coupling transformer T 2  together with a transformer T 2  primary coil Ld and a transformer T 2  primary coil Lb and receives AC power by electromagnetic induction. The number of windings of the exciting coil L 1  is 1, diameter of the wire of the exciting coil L 1  is 3 mm, and diameter of the exciting coil L 1  itself is 210 mm. Current I 1  flowing in the exciting circuit  1110  is AC. The direction of an arrow in the diagram of the exciting circuit  1110  indicates the positive direction, and direction opposite to the direction of the arrow indicates the negative direction. 
     The feeding coil circuit  1120  has the same configuration as that of the feeding coil circuit  1120  described in the third embodiment and resonates at the resonance frequency fr=100 kHz. The receiving coil circuit  1130  and loading circuit  1140  have the same configurations as those of the second and third embodiments. 
     The power circuit  1206  is a push-pull circuit operating at a drive frequency fo and has a vertically symmetrical configuration as illustrated in  FIG. 27 . The exciting circuit  1110  receives AC power at the drive frequency fo from the power circuit  1206 . In this case, the currents I 1  to I 4  at the drive frequency fo flow in the exciting circuit  1110 , feeding coil circuit  1120 , receiving coil circuit  1130 , and loading circuit  1140 . When the drive frequency fo and resonance frequency fr coincide with each other, that is, when the drive frequency fo assumes 100 kHz, the feeding coil circuit  1120  and receiving coil circuit  1130  magnetically resonate, maximizing the power transmission efficiency. 
     An oscillator  1202  is connected to the primary side of a gate-drive transformer T 1  included in the power circuit  1206 . The oscillator  1202  generates AC voltage at the drive frequency fo. The AC voltage causes current to flow in a transformer T 1  primary coil Lh alternately in both positive and negative directions. The transformer T 1  primary coil Lh, transformer T 1  secondary coil Lg, and transformer T 1  secondary coil Lf constitute a gate-drive coupling transformer T 1 . Electromagnetic induction causes current to flow also in the transformer T 1  secondary coil Lg and transformer T 1  secondary coil Lf alternately in both positive and negative directions. 
     The secondary coil of the transformer T 1  is center-point grounded. That is, one ends of the transformer T 1  secondary coil Lf and transformer T 1  secondary coil Lg are connected to each other and directly grounded. The other end of the transformer T 1  secondary coil Lf is connected to the gate of a switching transistor Q 1 , and the other end of the transformer T 1  secondary coil Lg is connected to the gate of a switching transistor Q 2 . The source of the switching transistor Q 1  and source of the switching transistor Q 2  are also grounded. Thus, when the oscillator  1202  generates AC voltage of the drive frequency fo, voltage Vx (Vx&gt;0) of the drive frequency fo is applied alternately to the gates of the switching transistors Q 1  and Q 2 . As a result, the switching transistors Q 1  and Q 2  are alternately turned on/off at the drive frequency fo. 
     The drain of the switching transistor Q 1  is connected in series to a transformer T 2  primary coil Ld. Similarly, the drain of the switching transistor Q 2  is connected in series to a transformer T 2  primary coil Lb. A smoothing inductor La and a power supply Vdd are connected to the connection point between the transformer T 2  primary coil Ld and transformer T 2  primary coil Lc. Further, a capacitor CQ 1  is connected in parallel to the source-drain of the switching transistor Q 1 , and a capacitor CQ 2  is connected in parallel to the source-drain of the switching transistor Q 2 . 
     The capacitor CQ 1  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 1 , and capacitor CQ 2  is inserted so as to shape the voltage waveform of the source-drain voltage VDS 2 . Even if the capacitors CQ 1  and CQ 2  are omitted, the wireless power feeding using the power circuit  1206  can be achieved. In particular, in the case where the drive frequency fo is low, it is easily possible to maintain the power transmission efficiency even if the capacitors are omitted. 
     The input impedance of the exciting circuit  1110  is 50 (Ω). The number of windings of the transformer T 2  primary coil Lb and the number of windings of the transformer T 2  primary coil Ld are set such that the output impedance of the power circuit  1206  is equal to the input impedance of 50(Ω). When the output impedance of the power circuit  1206  and input impedance of the exciting circuit  1110  coincide with each other, the power circuit  1206  has the maximum output. 
     When the switching transistor Q 1  is turned conductive (ON), the switching transistor Q 2  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “first current path  1112 ”) at this time is from the power supply Vdd through the smoothing inductor La, transformer T 2  primary coil Ld, and switching transistor Q 1  to the ground. The switching transistor Q 1  functions as a switch for controlling conduction/non-conduction of the first current path  1112 . 
     When the switching transistor Q 2  is turned conductive (ON), the switching transistor Q 1  is turned non-conductive (OFF). A main current path (hereinafter, referred to as “second current path  1114 ”) at this time is from the power supply Vdd through the smoothing inductor La, transformer T 2  primary coil Lb, and switching transistor Q 2  to the ground. The switching transistor Q 2  functions as a switch for controlling conduction/non-conduction of the second current path  1114 . 
     Also in the wireless power transmission system  1108 , the resistors R 1  and R 2  are connected to both ends of the oscillator  1202 , and the voltage phase is measured from the potential Vp 1  at the connection point A between the resistors R 1  and R 2 . In the fourth embodiment, the detection coil LSS is provided near the exciting circuit  1110 , and a part of the exciting circuit  1110  and detection coil LSS constitute a coupling transformer. Inductive current ISS is made to flow in the detection coil LSS by a magnetic field generated by AC current I 1 . The current phase is measured based on the inductive current ISS according to the same method as the first embodiment or second embodiment. The phase difference td between the current phase and voltage phase is detected by the phase detection circuit  1150 , and the drive frequency tracking circuit  1152  adjusts the drive frequency fo of the oscillator  1202 , thereby maintaining the resonance state. 
       FIG. 28  is a system configuration view of the wireless power transmission system  1108  according to a first modification of the fourth embodiment. Components designated by the same reference numerals as those of  FIG. 27  have the same or corresponding functions as those in  FIG. 27 . In the system configuration of  FIG. 27 , the exciting circuit  1110  and detection coil LSS constitute a coupling transformer by sharing the core  1154 . However, in the system configuration of  FIG. 28 , the feeding coil circuit  1120  and detection coil LSS constitute a coupling transformer by sharing the core  1154 . 
     It is possible to use not only the exciting circuit  1110  or feeding coil circuit  1120  but also the receiving coil circuit  1130  or loading circuit  1140  as the primary coil to constitute a coupling transformer so as to cause the detection coil LSS to generate the inductive current ISS. The inductive current ISS may be generated using the detection coil circuit  1170  described with reference to  FIGS. 24 and 25 . 
       FIG. 29  is a system configuration view of the wireless power transmission system  1108  according to a second modification of the fourth embodiment. Components designated by the same reference numerals as those of  FIGS. 27 and 28  have the same or corresponding functions as those in  FIGS. 27 and 28 . In the wireless power transmission system  1108  according to the second modification, the power circuit  1206  directly drives the feeding coil circuit  1120  without intervention of the exciting circuit  1110 . 
     The feeding coil circuit  1120  of the wireless power transmission system  1108  is a circuit in which the transformer T 2  secondary coil Li is connected in series to the feeding coil L 2  and capacitor C 2 . The transformer T 2  secondary coil Li constitutes the coupling transformer T 2  together with the transformer T 2  primary coil Lb and transformer T 2  primary coil Ld and receives AC power from the power circuit  1206  by electromagnetic induction. As described above, the AC power may be directly fed from the power circuit  1206  to the feeding coil circuit  1120  without intervention of the exciting circuit  1110 . 
     The wireless power transmission systems  300 ,  1100 ,  1106 , and  1108  have been described based on the respective embodiments. In the first embodiment, the exciting circuit  110 , feeding coil circuit  120 , receiving coil circuit  130 , and loading circuit  140  resonate at the same resonance frequency fr, so that if some load is added to these circuits, the Q-value reacts with high sensitivity. In the case of the wireless power transmission system  300  of the first embodiment, the current I 2  flowing in the feeding coil L 2  is not set as a measurement target, but current passing through the switching transistor Q 2  included in the power circuit  200  is set as the measurement target, making it easy to suppress the influence on the Q-value of the feeding coil circuit  120 . That is, it is possible to always monitor whether the drive frequency fo and resonance frequency fr coincide with each other while suppressing influence on the system&#39;s resonance characteristics caused by the measurement procedure. 
     As described with reference to  FIG. 5 , etc., in the case of wireless power feeding of the magnetic field resonance type, the coincidence degree between the resonance frequency fr and drive frequency fo gives great influence on the power transmission efficiency. Providing the phase detection circuit  150  or drive frequency tracking circuit  152  allows the drive frequency fo to automatically track a change of the resonance frequency fr, making it easy to maintain the power transmission efficiency at its maximum value even if use conditions are changed. 
     In the case where the wireless power transmission system  300  according to the first embodiment was used to perform an experiment under a condition that the distance between the exciting coil L 1  and feeding coil L 2  is made to equal to the diameters of the feeding coil L 2  and receiving coil L 3 , about 70% of the power transmitted from the feeding coil circuit  120  could be taken from the loading circuit  140 . 
     Also in the second to fourth embodiments, the feeding coil L 2 , receiving coil L 3 , and loading coil L 4  resonate at the same resonance frequency fr, so that if some load is connected to these coils, the Q-value reacts with high sensitivity. The same can be said in the case where the exciting coil L 1  is used. In the second to fourth embodiments, the AC power itself to be transmitted/received is not set as a measurement target, but the inductive current ISS is generated by the AC magnetic field generated at the transmission/reception time of the AC power so as to measure the current phase. Therefore, it is easily suppress the influence of the measurement procedure on the system&#39;s resonance characteristics (Q-value). 
     Also in the second to fourth embodiments, providing the phase detection circuit  1150  or drive frequency tracking circuit  1152  allows the drive frequency fo to automatically track a change of the resonance frequency fr, making it easy to maintain the power transmission efficiency at its maximum value even if use conditions are changed. 
     The above embodiments are merely illustrative of the present invention and it will be appreciated by those skilled in the art that various modifications may be made to the components of the present invention and a combination of processing processes and that the modifications are included in the present invention.