Patent Publication Number: US-9407178-B2

Title: Method and apparatus for controlling an electric machine in a six-step mode

Description:
TECHNICAL FIELD 
     The present disclosure generally relates to controlling alternating current (AC) motor/generators, and more particularly relates to apparatus, systems and methods for controlling AC motor/generators. 
     BACKGROUND 
     Synchronous frame current regulators are employed for current control of AC motor/generators, such as three-phase permanent magnet synchronous electric motors (electric machines). By providing dynamic control over a wide frequency range, synchronous frame current regulators are suited to many industrial applications. 
     Control of AC motor/generators, such as three-phase permanent magnet synchronous electric motors (electric machines) is accomplished using a three-phase pulsewidth-modulated (PWM) inverter. A PWM inverter can be controlled in several different operation modes, including, e.g., a space vector PWM (SVPWM) mode and a six-step mode. Output voltage magnitude of the inverter at the fundamental frequency becomes its maximum only when an inverter operates in the six-step mode. Due to this voltage magnitude characteristic, operation in the six-step mode can increase torque capability of an electric machine compared to known SVPWM operation or discontinuous space vector PWM (DPWM) operation in the field-weakening region where the voltage magnitude is the major limiting factor of the torque capability. However, voltage magnitude is not controllable in the six-step mode. Only voltage angle can be adjusted in the six-step mode. This is equivalent to loss of 1 Degree-Of-Freedom (DOF) in controllability compared to operation in the normal SVPWM mode or the DPWM mode. Because of this DOF loss, it has proven challenging to employ an asynchronous frame current regulator with a PWM inverter operating in the six-step mode. 
     SUMMARY 
     A voltage source inverter controller for controlling an inverter electrically connected to a permanent magnet synchronous multi-phase AC electric machine includes a current command generator, a six-step flux controller and a current regulator. The six-step flux controller generates a flux modifier to regulate flux in a flux-weakening speed/load operating region of the electric machine when operating the electric machine in a six-step mode. The current command generator converts a desired torque to three-phase desired currents that are input to a dq0-dq transformer and combined with the flux modifier to determine a modified-flux direct-quadrature (dq) current request. The current regulator includes a proportional-integral feedback controller, anti-windup elements, a dq voltage limit element and a voltage magnitude limiter. The proportional-integral feedback controller and the anti-windup elements perform closed-loop current control on the modified-flux dq current request to determine commanded dq voltages. This includes the dq-voltage limit element and the voltage magnitude limiter imposing limits to the commanded dq voltages, and the inverter converting the limited commanded dq voltages to pulsewidth-modulated stator currents to drive the electric machine in the six-step mode. 
     The above features and advantages, and other features and advantages, of the present teachings are readily apparent from the following detailed description of some of the best modes and other embodiments for carrying out the present teachings, as defined in the appended claims, when taken in connection with the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       One or more embodiments will now be described, by way of example, with reference to the accompanying drawings, in which: 
         FIG. 1  schematically illustrates a voltage source inverter controller including a current regulator and a six-step flux controller for controlling an inverter electrically connected to a multi-phase AC electric motor/generator (electric machine) in a six-step mode, in accordance with the disclosure; 
         FIG. 2  schematically shows an embodiment of a current regulator that can be employed by a controller in a system for controlling an electric machine in a six-step mode, in accordance with the disclosure; and 
         FIG. 3  schematically shows an embodiment of a six-step inverter control routine for determining when to control operation of an embodiment of an electric machine in a six-step mode including an embodiment of a threshold torque calibration including a first speed/torque operating region wherein the six-step mode is prohibited and a second speed/torque operating region wherein the six-step mode is permitted, in accordance with the disclosure. 
     
    
    
     DETAILED DESCRIPTION 
     Referring now to the drawings, wherein the depictions are for the purpose of illustrating certain exemplary embodiments only and not for the purpose of limiting the same,  FIG. 1  schematically illustrates a Voltage Source Inverter (VSI) controller  100  for controlling an inverter  30  electrically connected to a multi-phase AC electric motor/generator (electric machine)  40  in accordance with the disclosure. The electric machine  40  is preferably a permanent magnet synchronous device including a stator and a rotor arranged in a star configuration, although the concepts described herein are not so limited. The VSI controller  100  selectively controls operation of the inverter  30  in one of a PWM mode and a six-step mode. 
     The VSI controller  100  controls torque output from the electric machine  40  through the inverter  30 , which electrically connects to a high-voltage DC electric power supply. Control methods for switching among inverter states to regulate torque output of the electric machine  40  include operating in either a PWM mode or a six-step mode. In the PWM mode, the inverter  30  switches rapidly among two of the non-zero states and one of the zero states. The VSI controller  100  specifies what fraction of the time should be spent in each of the three states by specifying PWM duty cycles. The VSI controller  100  updates the PWM duty cycles at regular intervals such that the frequency of updates is significantly higher than the frequency of the rotor rotation. In the six-step mode, the inverter  30  cycles through the six non-zero states once per cycle of the rotor of the electric machine  40  to produce an AC voltage and current in each winding of the stator. A rotor cycle is defined relative to motor poles and does not necessarily correspond to a complete revolution of the rotor. 
     The VSI controller  100  includes a current regulator  20  and a six-step flux controller  50  for controlling operation of inverter  30  to control operation of the electric machine  40  in the six-step mode. The amplitude of the AC voltage is dictated by the magnitude of DC voltage on the high-voltage DC bus that electrically connects a high-voltage electric power source to the inverter  30 . The torque is dictated by the DC voltage, the rotor speed, and the phase difference between these quasi-sinusoidal AC voltage signals and the rotor position, and is further controlled by operating the control system in six-step mode. The VSI controller  100  issues commands to the inverter  30  indicating when to switch to the next state in the sequence. 
     Inputs to the VSI controller  100  include a desired torque Te*  11  and a desired outer voltage |U dq *|  13 . A current command generator  10  converts the desired torque Te*  11  to three-phase (dq0) desired currents i dq0   r *  12 , which are input to a dq0-dq transformer  15 . The dq0-dq transformer  15  employs a flux modifier Δβ  52  to determine a modified-flux direct-quadrature (dq) current request i dq   r *  17 . The flux modifier Δβ  52  is determined by the six-step flux controller  50  as described herein. The dq0-dq transformer  15  combines the flux modifier Δβ  52  with a flux term β calculated based upon the desired torque Te*  11  and the rotational speed ω of the rotor. The dq0-dq transformer  15  recalculates the modified-flux direct-quadrature (dq) current request i dq   r *  17  employing the modified flux (β+Δβ), which includes converting the dq0 desired currents i dq0   r *  12  to the modified-flux dq current request i dq   _   r *new  17  employing known dq0-dq transform methodologies. A dq0-dq transform reduces three-phase AC quantities, e.g., u a , u b  and u c  into the dq components, e.g., u d  and u q  to facilitate filtering and control, with the active and reactive powers controlled independently by controlling the dq components. 
     The six-step flux controller  50  operates when operating the electric machine  40  in the six-step active mode. A process of determining when to control operation of an embodiment of the electric machine  40  in the six-step active mode is described with reference to  FIG. 3 . The six-step flux controller  50  operates as follows. The magnitude of the desired outer voltage |u dq *|  13  is reduced by a magnitude of the commanded outer voltage |u dq   _   out |  23  using a difference element  45  to determine an outer voltage error term  47  that is input to the six-step flux controller  50 . The six-step flux controller  50  applies a proportional gain k p    66  to the outer voltage error term  47  and applies an integral gain k i    63  that is subject to a time delay  64  and upper and lower integral boundaries  65  to the outer voltage error term  47 . The resultants are added by a summing element  67 , and subjected to upper and lower flux boundaries  68  and a directional sign  69 , wherein negative flux (−) is associated with negative torque, i.e., operation of the electric machine  40  in an electric power regeneration mode and positive flux (+) is associated with positive torque, i.e., operation of the electric machine  40  in a torque generation mode to determine the flux modifier Δβ  52  for the dq current command. The six-step flux controller  50  increases flux in a flux-weakening speed/load operating region of the electric machine  40 . The current command and current command angle, β, changes in a clockwise direction. The actual dq current follows its command until magnitude of the commanded outer voltage, |u dq |  23  is less than six-step, 3/π≈0.955, and once after |u dq | reaches the full six-step, the actual dq current moves along with an ellipse that is defined by full six-step voltage magnitude. However, the actual current may have out-and-into movement when the magnitude of the commanded outer voltage |u dq | reaches its full six-step magnitude. This kind of current movement can be critical if the motor speed is in medium range so the controller is in the six-step active mode when current command is at a maximum. The magnitude of the actual current can be larger than its command, and that can damage the inverter and motor. Thus, voltage clamping is employed as described with reference to  FIG. 2 . 
     The six-step flux controller  50  supplants a flux-weakening controller when operating the inverter  30  and the electric machine  40  in the six-step active mode. A difference between magnitudes of the outer voltage |u dq | and desired outer voltage |u dq   _   out | changes the angle of the dq current command and corresponding flux β. The command of the outer voltage magnitude |u dq |* is determined based upon the six-step voltages with a recommended calibration factor of 1.2 in one embodiment. 
     The six-step flux controller  50  prevents jittering of a PWM alignment flag because it keeps the magnitude of the outer voltages greater than the six-step limit. This feature is helpful especially when the motor temperature is high thus weakening the magnet flux. The six-step flux controller  50  helps to keep the magnitude of the actual current as close as possible to that of the command current, because the outer voltage magnitude is equivalent to the current error. In other words, the six-step flux controller automatically finds an operating point that is available with the full six-step voltages and the magnitude of the current command. 
     The modified-flux dq current request i dq   _   r *new  17  is reduced by a current feedback term i dq   _   r *fdbk  37  using a difference element  18 . The current feedback term i dq   _   r *fdbk  37  is derived from monitored current commands u abc    35  between the inverter  30  and the electric machine  40 . A current regulator input term  19  including a difference between the modified-flux dq current request i dq   _   r *new  17  and the dq current feedback term i dq   _   r *fdbk  37  is input to the current regulator  20 , which operates as described with reference to  FIG. 2  to generate the dq voltage commands u dq    22 . The current regulator input term  19  includes a d-axis synchronous frame stator current I ds   r    51  and a q-axis synchronous frame stator current I qs   r    53 , which are shown and described with reference to  FIG. 2 . 
     The dq voltage commands u dq    22  are provided as inputs to the inverter  30 , which is preferably a pulsewidth modulation (PWM) voltage source inverter that electrically couples to the AC motor  40 . In response to the dq voltage commands u dq    22 , the inverter  30  produces AC current commands u abc    35  that generate stator current in the windings of the electric machine  40  to drive rotation and torque output from the electric machine  40 . Reverse transformation module  36  converts the AC current commands u abc    35  to the current feedback term dq current feedback i dq   _   r *fdbk  37  using known dq-abc conversion methods. 
     The reverse transformation module  36  transforms the AC current commands u abc    35 , e.g., three-phase sinusoidal stator currents i as , i bs  and i cs  into direct-quadrature (dq) terms including the current feedback i dq   _   r *fdbk  37 , which includes a d-axis synchronous frame commanded current I ds   r *  51  and a q-axis synchronous frame commanded current I qs   r *  53 , which correspond to d-axis synchronous frame commanded current I ds   r *  201  and a q-axis synchronous frame commanded current I qs   r *  203  shown and described with reference to  FIG. 2 . 
     In one embodiment, sensing devices may be coupled to the electric machine  40  to sample the AC signals and supply these and other measured quantities to the controller  10 . Measured quantities can include supply potential, e.g., a battery potential or high-voltage DC bus voltage v dc  and the three-phase sinusoidal stator currents i as , i bs  and i cs , although measurement of two of the phase currents may be sufficient when the electric machine  40  is a Y-connected machine without a neutral line. Rotational speed ω of the electric machine  40  and a rotor phase angle θ r  of the electric machine  40  are monitored, preferably with a sensor  41 , which can be any suitable rotational speed/position sensor such as a resolver or a Hall-effect sensor. 
     The VSI controller  100  executes one or more programs to optimize commanded currents for a predetermined control parameter to determine operating inputs in the form of modified commanded currents, commanded voltages, torque commands, or the like to control the electric machine  40  via the current regulator  20 . One or more of the components of the VSI controller  100  may be embodied in software or firmware, hardware, such as an application specific integrated circuit (ASIC), an electronic circuit, a processor (shared, dedicated, or group) and memory that execute one or more software or firmware programs, a combinational logic circuit, and/or other suitable components, or a combination thereof. In one embodiment, the VSI controller  100  is partitioned into one or more processing modules that are associated with one or more of the controller operations. For example, the current regulator  20  may be implemented as one of these processing modules. Although not shown, the controller  10  may include additional modules, such as a commanded current source, a torque module and a field-weakening voltage control module. 
     The terms controller, control module, module, control, control unit, processor and similar terms refer to any one or various combinations of Application Specific Integrated Circuit(s) (ASIC), electronic circuit(s), central processing unit(s), e.g., microprocessor(s) and associated memory and storage devices (read only, programmable read only, random access, hard drive, etc.) executing one or more software or firmware programs or routines, combinational logic circuit(s), input/output circuit(s) and devices, signal conditioning and buffer circuitry and other components to provide a described functionality. Software, firmware, programs, instructions, control routines, code, algorithms and similar terms mean any controller-executable instruction sets including calibrations and look-up tables. Each controller executes control routine(s) to provide desired functions, including monitoring inputs from sensing devices and other networked controllers and executing control and diagnostic routines to control operation of actuators. Routines may be executed at regular intervals, for example each 100 microseconds. Communications between controllers and between controllers, actuators and/or sensors may be accomplished using a direct wired link, a networked communications bus link, a wireless link or any another suitable communications link. 
       FIG. 2  schematically shows an embodiment of the current regulator  20  that can be employed by a controller in a system for controlling an AC electric motor/generator, e.g., the VSI controller  100  described with reference to  FIG. 1 . The current regulator  20  generates commanded direct and quadrature voltages V ds   r *  281  and V qs   r *  283 , respectively, which can be converted to commanded six-step direct and quadrature voltages V ds   r    211  and V qs   r    213 , respectively, via a voltage magnitude limiter  290 . The inverter  30  converts the commanded direct and quadrature voltages V ds   r *  281  and V qs   r *  283  and the commanded six-step direct and quadrature voltages V ds   r    211  and V qs   r    213  to pulsewidth-modulated stator currents i as , i bs  and i cs  to drive an electric motor, e.g., the electric machine  40  described with reference to  FIG. 1 . 
     The current regulator  20  is a complex PI controller that includes a dq-voltage limit element  280  and anti-windup elements that include a current command compensation, thus providing stability in a heavy wind-up condition. Inputs to the current regulator  20  include command inputs including a d-axis synchronous frame commanded current I ds   r *  201  and a q-axis synchronous frame commanded current I qs   r *  203 . Feedback inputs to the current regulator  20  include the d-axis synchronous frame stator current I ds   r    51  and the q-axis synchronous frame stator current I qs   r    53 . 
     An anti-windup scheme limits operation in the six-step PWM mode as follows. Difference blocks  247  and  277  each calculate a difference between the commanded six-step direct and quadrature voltages V ds   r    211  and V qs   r    213 , respectively, and the corresponding commanded direct and quadrature voltages V ds   r *  281  and V qs   r *  283 , respectively. The resultants are multiplied by one of gains k ad    241  and k qd    271 , respectively, and multiplied by one of second gains kdd  243  and k q    273 , respectively, for addition to the d-axis synchronous frame commanded current I ds   r *  201  and the q-axis synchronous frame stator current I qs   r    53 , respectively, in summing blocks  221  and  251 , respectively. The second gains k d    243  and k q    273 , respectively, provide current command compensation for anti-windup. Difference blocks  223  and  253  calculate a difference between the resultants and the d-axis synchronous frame commanded current I ds   r    51  and the q-axis synchronous frame stator current I qs   r    53 , respectively. 
     The calculated differences from difference blocks  223 ,  253  are subjected to complex proportional-integral controls that include cross-over feedback control parameters. The complex proportional-integral controls include proportional gains k pd    229  and k pq    259 , integral difference elements  225  and  255 , delays  227  and  257 , multipliers  230  and  260 , cross-over integrator gains k′ id  and k′ iq    245  and  275 , integrator gains k id  and k iq    231  and  261 , summing elements  233 ,  237 ,  263  and  267 , integrator clamp  287 , gains R d    239  and R q    269 , and summing elements  235  and  265 , preferably arranged as shown with reference to  FIG. 2 . The outputs of the summing elements  235  and  265  are input to the dq-voltage limit element  280 , which clamps the output voltage based upon a maximum voltage V lim  and produces the DC commanded voltages V ds   r *  281  and V qs   r *  283 , respectively. The DC commanded voltages V ds   r *  281  and V qs   r *  283 , respectively are input to the voltage magnitude limiter  290  that calculates the commanded six-step direct and quadrature voltages V ds   r    211  and V qs   r    213  for controlling the electric machine  40 . 
     Anti-windup is accomplished by multiplying gains k ad    241  and k qd    271 , respectively, and second gains k d    243  and k q    273 , respectively, for addition to the d-axis synchronous frame commanded current I ds   r *  201  and the q-axis synchronous frame stator current I ds   r    203 , respectively, in summing blocks  221  and  251 , respectively. The anti-windup compensation operates as follows. When the steady-state dq current command is near the voltage limit, the anti-windup algorithm can create steady-state error in the actual dq current. The amount of this steady-state current error is same as the voltage error used by the anti-windup elements, and is determined as follows. 
     
       
         
           
             
               
                 
                   
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     Using the above equation, the current command itself can be compensated as follows to eliminate the steady-state error. 
     
       
         
           
             
               
                 
                   
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     To give some degree of freedom in calibration process, two additional gains are created to adjust the amount of the current compensation. The final equation is as follows. 
     
       
         
           
             
               
                 
                   
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     The outer circle limit of dq voltage commands including integrator clamping is depicted as v lim    282  in the dq-voltage limit element  280 . 
     In the six-step active mode, the current commands are determined with a voltage limit to prevent jittering between the six-step mode and the PWM mode. However, this can cause steady-state error in the dq current that can be accumulated in the integrator elements. To prevent integrator saturation, an additional voltage limit is employed. Thus, the current controller includes the dq-voltage limit element  280  and the voltage magnitude limiter  290 . When the output of the current controller is clamped with dq-voltage limit element  280 , the integrators are multiplied by 
                   V   lim            v   dq     r   *              ⁢   285     ,         
causing the integrators to be clamped with phase information preserved.
 
     This outer voltage also can be used as an indicator of the current error in the six-step active mode, because the magnitude of the inverter output voltage is fixed to six-step, 
               2   π     *   V   ⁢           ⁢   d   ⁢           ⁢   c         
in me six-step active mode. In other words, as this voltage grows higher, the actual dq current has more steady-state error compared to its command. For this reason, this outer voltage is consumed in the six-step flux controller  50 .
 
     Complex PI control is accomplished by applying the cross-over integrator gains k′ id  and k′ iq    245  and  275  into opposite q, d integrators at summing elements  267 ,  237  respectively. 
     Feedback control is accomplished by first voltage limit  280  that generates the voltage 
                 V   lim            v   dq     r   *              ⁢   285         
that is clamped by integrator clamp  287  and inserted into the integrators at both the multipliers  230  and  260 .
 
     Operation in the six-step mode is limited by the voltage magnitude limiter  290 . 
     The complex PI controller with anti-windup including the current error is integrated with anti-windup terms first, and then the integrated current error is branched out for cross-coupling. Voltage clamping is thus applied to the actual dq currents. 
     In operation, the controller  10  may retrieve the commanded currents from a commanded current table stored in a memory of the controller  10 . The commanded current table is preferably optimized for one or more predetermined control parameters (e.g., system efficiency) and may be derived from any number of models for optimizing the desired control parameter(s). Additionally, the commanded current table may be predetermined based on voltage and current limits of the electric machine  40  such that the commanded current source applies an appropriate amount of d-axis and q-axis currents to the electric machine  40  to produce a desired torque (e.g., with high efficiency) and maintain current regulation stability. The inverter voltage limits may be predetermined based on the supply voltage. 
       FIG. 3  schematically shows an embodiment of a six-step activation routine  300  for determining when to control operation of an embodiment of the electric machine  40  described herein in a six-step active mode. Table 1 is provided as a key wherein the numerically labeled blocks and the corresponding functions are set forth as follows, corresponding to the six-step active routine  300 . 
     
       
         
           
               
               
             
               
                 TABLE 1 
               
               
                   
               
               
                 BLOCK 
                 BLOCK CONTENTS 
               
               
                   
               
             
            
               
                 302 
                 Start 
               
               
                 304 
                 Select six-step threshold torque 
               
               
                 306 
                 Is commanded torque greater than  
               
               
                   
                 threshold torque? 
               
               
                 308 
                 Permit six-step active routine 
               
               
                 310 
                 Prohibit six-step active routine 
               
               
                 312 
                 Index 
               
               
                 314 
                 End 
               
               
                   
               
            
           
         
       
     
     The six-step activation routine  300  is a scheduled task that executes periodically during ongoing operation, e.g., once every 100 microseconds or once every 500 microseconds. Upon initiating the six-step active routine  300  ( 302 ), a six-step threshold torque is selected based upon motor speed ( 304 ). A threshold torque calibration table is shown graphically, including motor speed on the horizontal axis  330  in relation to commanded motor torque on the vertical axis  320 . Area  323  indicates a speed/torque operating region wherein the six-step active routine is prohibited and area  325  indicates a speed/load operating region wherein the six-step active routine is permitted, and further relates to flux-weakening speed/load operating region of the electric machine  40 . Line  324  including minimum speed  334  delineates a threshold torque between area  323  and area  325  that is associated with increasing commanded motor torque. Line  322  including minimum speed  332  delineates a threshold torque between area  323  and area  325  that is associated with decreasing commanded motor torque. Torque associated with line  324  is greater than torque associated with line  322  throughout the region associated permitting the six-step active routine. Such speed/torque delineation between areas  323  and  325  allow introduction of hysteresis into the six-step active routine  300 . The threshold torque calibration table can be implemented in software as a searchable multi-dimensional table, equations, or in any other suitable executable form. 
     When the commanded torque is greater than the threshold torque for the commanded torque as determined with reference to the threshold torque calibration table ( 306 )( 1 ), the six-step active control routine is permitted ( 308 ) and the VSI controller  100  described with reference to  FIGS. 1 and 2  is employed to control the electric machine  40  using six-step inverter control. 
     When the commanded torque is less than the threshold torque for the commanded torque as determined with reference to the threshold torque calibration table ( 306 )( 0 ), the six-step active control routine is prohibited ( 310 ) and control of the electric machine  40  is accomplished using PWM inverter control. The iteration is indexed ( 312 ), and ends ( 314 ). 
     The VSI controller  100  described herein provides a modified current regulator in combination with a six-step flux controller and a six-step activation routine to provide closed loop current control when controlling an electric machine in six-step mode. Transitions between the six-step mode and the PWM mode can be executed without employing transient response management circuitry or algorithms that would otherwise be required to minimize current and torque spikes. 
     The detailed description and the drawings or figures are supportive and descriptive of the present teachings, but the scope of the present teachings is defined solely by the claims. While some of the best modes and other embodiments for carrying out the present teachings have been described in detail, various alternative designs and embodiments exist for practicing the present teachings defined in the appended claims.