Patent Publication Number: US-6985099-B1

Title: Automatic gain control with digital filtering for radio-frequency communications systems

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates generally to automatic gain control (AGC) for use in radio-frequency (RF) communications systems. More specifically, but without limitation thereto, the present invention relates to an automatic gain control with digital filtering for mitigating dynamic range reduction due to out-of-band interference. 
     Typical radio frequency communications signals are divided into a number of channels within a signal band centered around a carrier frequency. For example, 10 channels each 5 megahertz (MHz) wide can fit into a 50 MHz wide frequency band centered at a carrier frequency of, for example, 2 gigahertz (GHz). About 99% of the signal in each channel is contained in a band of frequencies about 4.4 MHz wide, ordinarily leaving a gap between adjacent channels to avoid mutual interference. During transmission of the signal, however, signal from one channel may cross over into other channels (or bands) causing out-of-band interference. Much time, effort, and ingenuity has been devoted to the problem of filtering out this out-of-band interference. 
     One of the difficulties encountered in signal filtering is that radio frequency signals in the gigahertz range are much more difficult to filter than signals at lower frequencies. One approach to solving this filtering problem is to translate the radio frequency signal to an intermediate frequency (IF) signal. For example, if a 5 MHz channel is transmitted on a carrier in the gigahertz range from 1.995 GHz to 2.000 GHz, the carrier may be translated to an intermediate frequency at some lower frequency, for example, 700 MHz. The 5 MHZ channel would then occupy from 695 MHz to 700 MHz. At this lower frequency, filtering is relatively straightforward. 
     Analog filters in the radio frequency, intermediate frequency, and baseband ranges are typically used to mitigate out-of-band interference. Disadvantageously, analog filters can be expensive, especially for the higher frequencies. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other aspects, features and advantages of the present invention will be more apparent from the following more specific description thereof, presented in conjunction with the following drawings wherein: 
         FIG. 1  is a block diagram of a typical radio frequency communications receiver using a conventional automatic gain control; 
         FIG. 2  is a block diagram of a radio frequency communications receiver using digitally filtered automatic gain control according to an embodiment of the present invention; 
         FIG. 3  is a block diagram of an infinite impulse response filter for the automatic gain control of  FIG. 2 ; 
         FIG. 4  is a graph of the magnitude response versus frequency of the infinite impulse response filter of  FIG. 3 ; 
         FIG. 5  is a graph of the phase response versus frequency of the infinite impulse response filter of  FIG. 3 ; and 
         FIG. 6  illustrates frequency response comparisons as between an “deal” IF filter and a “practical” IF filter. 
     
    
    
     Corresponding reference characters indicate corresponding components throughout the several views of the drawings. 
     DETAILED DESCRIPTION OF THE DRAWINGS 
     The present invention advantageously addresses the needs above as well as other needs by providing a method and apparatus for digitally filtered automatic gain control. 
     In one embodiment, the invention may be characterized as an automatic gain control that includes a digital lowpass filter for filtering a series of digital samples generated by an analog-to-digital converter to generate a lowpass filtered digital sample series; a power averager coupled to the digital lowpass filter for calculating an average power of the lowpass filtered digital sample series; and a lookup table coupled to the power averager for setting the selectable gain of an amplifier coupled to the analog-to-digital converter as a function of the average power. 
       FIG. 1  is a block diagram of a typical radio frequency communications receiver  100  using a conventional automatic gain control (AGC). Shown are an antenna  102 , a radio frequency filter circuit  104 , an intermediate frequency converter  106 , an intermediate frequency (IF) filter  108 A, herein provided as an “ideal” IF filter, an intermediate frequency amplifier  110 , an automatic gain control (AGC) circuit  112 , a demodulator  114 , an analog-to-digital (A/D) converter  118 , and a digital receiver  120 . 
     The antenna  102  receives as input a radio frequency signal containing the signal channels. The radio frequency signal may be transmitted from any source of radio frequency signals, for example, a satellite antenna. The radio frequency filter  104  removes a portion of out-of-band interference from the received radio frequency signal. The intermediate frequency converter  106  converts or translates the relatively high frequency radio frequency signal to a more readily filtered, lower intermediate frequency signal. The intermediate frequency filter  108 A removes harmonics from the intermediate frequency converter  106  and more of the out-of-band interference from the intermediate frequency signal. The intermediate frequency amplifier  110  amplifies the filtered intermediate frequency signal. The demodulator  114  translates the amplified intermediate frequency signal to a complex baseband signal. The complex baseband signal is illustrated by double lines, one for the in-phase or real component, and the other for the quadrature-phase or imaginary component. 
     The analog-to-digital converter  118  samples the complex baseband signal at a rate greater than the Nyquist rate to avoid aliasing and converts the complex baseband signal to complex digital samples, i.e., two separate series of digital samples that are representative of the real (in-phase) and imaginary (quadrature-phase) components of the communications signal respectively. The two series of digital samples output by the analog-to-digital converter  118  are then received as input by the digital receiver  120 . 
     The conventional automatic gain control circuit  112  estimates the power of the communications signal from the digital samples and periodically adjusts the gain of the intermediate frequency amplifier  110  so that the amplified intermediate frequency signal is scaled to the dynamic range of the analog-to-digital converter  118 . The purpose of scaling the intermediate frequency signal to the dynamic range of the analog-to-digital converter  118  is to maximize the signal-to-noise ratio at the input of the digital receiver  120 . A disadvantage of this approach is that out-of-band interference, if not adequately mitigated by the analog filters  104 ,  108 , and  116 , may reduce the available gain of the intermediate frequency amplifier  110 . The available gain of the intermediate frequency amplifier  110  is reduced when the out-of-band interference generates peak amplitudes that must be accommodated by the dynamic range of the analog-to-digital converter to avoid clipping, leaving fewer bits of dynamic range for the communications signal. The reduced gain of the intermediate frequency amplifier  110  reduces the dynamic range of the communications signal at the analog-to-digital converter  118  and correspondingly the signal-to-noise ratio (SNR) at the input of the digital receiver  120 . Out-of-band interference is especially common in multiple access systems because adjacent frequency bands are used concurrently. A solution to the problem of out-of-band interference is to improve the analog filters, but high-performance analog filters may be prohibitively expensive. 
       FIG. 2  is a block diagram of a radio frequency communications receiver  200  using digitally filtered automatic gain control. By introducing digital filtering into the automatic gain control, the requirements for the analog filters may be relaxed, substantially lowering the cost of the radio frequency communications receiver  200 . Alternatively, the out-of-band interference may be further reduced for the same analog filters, further increasing the signal-to-noise ratio at the digital receiver  120 . 
     Shown in  FIG. 2  are an antenna  102 , a radio frequency filter circuit  104 , an intermediate frequency converter  106 , an intermediate frequency (IF) filter  108 B, herein a “practical” IF filter embodiment, an intermediate frequency amplifier  110 , a filtered automatic gain control  202 , a demodulator  114 , an analog-to-digital (A/D) converter  118 , a digital receiver  120 , and a finite impulse response lowpass filter  122 . 
     The automatic gain control  202  includes a first decimator  204 , a digital infinite impulse response filter  206 , a second decimator  208 , an average power estimator  210 , and a gain look-up table  212 . 
     In operation, the radio frequency communications receiver  200  is similar to that of  FIG. 1  except as follows. The analog-to-digital converter  118  samples the filtered complex baseband signal to provide an adequate sampling output for the digital finite impulse response filter  122  at a sampling rate of, for example, four times the Nyquist sampling rate. The analog-to-digital converter  118  digitizes the complex baseband signal to a series of complex digital samples, i.e., two separate series of digital samples that are representative of the real (in-phase) and imaginary (quadrature-phase) components of the communications signal respectively. The two series of digital samples output by the analog-to-digital converter  118  are then received as input by the digital finite impulse response lowpass filter  122 . The digital finite impulse response lowpass filter  122  further removes out-of-band interference and outputs a digital output signal to the digital receiver  120 . The digital finite impulse response lowpass filter  122  and the digital receiver  120  may be implemented in hardware or software according to well known techniques. 
     The filtered automatic gain control  202  receives as input the two series of digital samples from the analog-to-digital converter  118 . Assuming that the front-end analog filters have sufficiently removed interference at frequencies higher than the Nyquist rate, the two series of digital samples may be decimated by a factor of two by the first decimator  204  without risk of aliasing. By reducing the sample rate by half, the first decimator  204  relaxes the performance required of the digital infinite impulse response filter  206 . 
     The digital infinite impulse response filter  206  attenuates out-of-band interference between half the Nyquist rate and the Nyquist rate from the decimated digital samples output by the first decimator  204 . The digital infinite impulse response filter  206  in this example has the desirable characteristics of simple implementation and minimal delay, however other types of filters may be used to remove frequencies between half the Nyquist rate and the Nyquist rate to suit specific applications. 
     The second decimator  208  again reduces the sample rate by half to relax the performance requirements of the average power estimator  210 . The average power estimator calculates an average power estimate as a running average of the signal power, for example, by calculating a sum of the squares of the two series of digital samples received from the second decimator  208 . The average power estimate is generated as output to the lookup table  212 . 
     The lookup table  212  contains amplifier gain coefficients for each average power estimate. The amplifier gain coefficients are precalculated as a function of the average power estimate, for example, the scalar factor between each average power estimate and the desired average output power within the dynamic range of the analog-to-digital converter  118 . Each amplifier gain coefficient adjusts the gain of the intermediate frequency amplifier  110  to minimize the difference between the corresponding average power estimate and the desired average output power within the dynamic range of the analog-to-digital converter  118 , thereby maintaining the average power of the amplified intermediate frequency signal at the full usable dynamic range of the analog-to-digital converter  118 . Because interference signal power may be higher than the communications signal power, some of the higher order bits of the analog-to-digital converter  118 , for example, the two most significant bits, are reserved as headroom to avoid overflow or clipping of the communications signal. Some of the lower order bits of the analog-to-digital converter  118 , for example, the two least significant bits, are used for precision in the digital infinite impulse response filter  206 . The remaining bits of the analog-to-digital converter  118 , for example, the middle four bits, are extracted by the digital finite impulse response lowpass filter  122  to filter the baseband signal for the digital receiver  120 . In this example, an eight-bit analog-to-digital converter may be used for the analog-to-digital converter  118 . 
       FIG. 3  is a block diagram of an infinite impulse response filter  206  for the automatic gain control of  FIG. 2  that can provide 20 dB attenuation between 0.8 of the Nyquist rate and 1.0 of the Nyquist rate. Identical infinite impulse response filters  206  may be used for each of the two series of digital samples. While the digital finite impulse response lowpass filter  122  is used for baseband filtering at the digital receiver  120  because of its low phase and magnitude distortion in the filtered digital baseband, the digital infinite impulse response lowpass filter  206  is a preferred choice for the filtered automatic gain control  202  because it is simple to implement and incurs minimal time delay in the signal. Also, the relatively greater phase distortion of an infinite impulse response (IIR) filter compared to a finite impulse response filter do not adversely affect the calculation of the average power by the average power estimator  210 . 
     By way of example, for code division multiple access (CDMA) applications requiring compliance with the CDMA2000 standard in which each channel has, for example, a bandwidth of 3.6864 MHz, the infinite impulse response filter  206  may be described by the following transfer function: 
               H   ⁡     (   z   )       =       0.3125   ⁢       (     1   +     z     -   1         )     2           (     1   +     0.5   ⁢   j   ⁢           ⁢     z     -   1           )     ⁢     (     1   -     0.5   ⁢   j   ⁢           ⁢     z     -   1           )                 (   1   )             
 
     The transfer function (1) may be implemented in either hardware or software as shown in  FIG. 3  by a first sum function  302 , a first sum register  304 , a first unit delay  306 , a third sum function  308 , a second unit delay  310 , a first multiplier  312 , a second multiplier  314 , a second sum function  316 , a second sum register  318 , a third sum register  320 , and a third multiplier  322 . 
     In operation, the output of the first decimator  204  is received as an N-bit wide input and summed by the first sum function  302 . The output of the sum function  302  is stored in the first sum register  304 . The first sum register  304  is N+1 bits wide, which is one bit wider than the output of the analog-to-digital converter  118 , to accommodate the output of the first sum function  302 . 
     The output of the first sum register  304  is delayed one sample period by the first unit delay  306  to generate a first delayed sum. 
     The first delayed sum output from the first unit delay  306  is multiplied by two by the first multiplier  312  and is delayed one sample period by the second unit delay  310  to generate a second delayed sum. The second delayed sum is multiplied by −0.25 by the second multiplier  314 . The output of the second multiplier  314  is summed by the first sum function  302 . The second delayed sum and the output of the first multiplier  312  are summed by the second sum function  316 . 
     The output of the second sum function  316  is stored in the second sum register  318 . The second sum register  318  is one bit wider than the output of the analog-to-digital converter  118  to accommodate the output of the second sum function  316 . The output of the second sum register  318  and the output of the first sum register  304  are summed by the third sum function  308 . The output of the third sum function  308  is stored in the third sum register  320 . The third sum register  320  is two bits wider than the output of the analog-to-digital converter  118  to accommodate the output of the third sum function  308 . 
     The output of the third sum register  320  is multiplied by 0.3125 by the third multiplier  322  to normalize the output of the third multiplier  322  to the digital sample series received as input by the first sum function  302 . The output of the third multiplier  322  has the same number of bits N as the analog-to-digital converter  118  and is the lowpass filtered output of the infinite impulse response lowpass filter  206 . 
       FIGS. 4 and 5  are graphs of the magnitude response and phase response versus frequency, respectively, of the infinite impulse response filter  206  shown in  FIG. 3 . Truncation effects at the output of the third multiplier  322  from normalizing the output to the input without rounding off the least significant bit are minimal when applied to an AGC circuit. Accordingly,  FIG. 6  is a graph showing the frequency response comparisons as between the “ideal” IF filter  108 A of  FIG. 1  and that of the “practical” IF filter  108 B frequency response of  FIG. 2 . A magnitude response of 10 dB or more is illustrated. 
     For a channel having a bandwidth of 3.68 MHz (i.e.; centered at zero extending to band edges at −1.84 MHz and +1.84 MHz) attenuation at the band edge (1.84 MHz) is 1.5 dB, while at 3.2 MHz attenuation is 26.5 dB. Because the filter passband is not entirely flat, the 8-bit output power is 0.046 dB above the unfiltered power for a code division multiple access (CDMA) signal. The lookup table  212  can be modified accordingly to compensate for this discrepancy. 
     Other modifications, variations, and arrangements of the present invention may be made in accordance with the above teachings other than as specifically described to practice the invention within the spirit and scope defined by the following claims.