Patent Publication Number: US-8994225-B2

Title: Wireless power transmission system and resonator for the system

Description:
CROSS REFERENCE TO RELATED APPLICATIONS: 
     This application is a U.S. national stage application under 35 USC 371 of International Application No. PCT/KR2010/004394 filed on Jul. 6, 2010, which claims the benefit of Korean Application No. KR 10-2009-0060984 filed on Jul. 6, 2009, in the Korean Intellectual Property Office, the entire disclosures of which are incorporated herein by reference for all purposes. 
     TECHNICAL FIELD 
     The following description relates to a wireless power transmission system, and more particularly, to a method for designing a resonator for a wireless power transmission system. 
     BACKGROUND ART 
     One of the wireless power transmission technologies may use a resonance characteristic of radio frequency (RF) devices. A resonator using a coil structure may require a change in a physical size based on a frequency. 
     DISCLOSURE OF INVENTION 
     In one general aspect, there is provided a wireless power resonator, including at least two unit resonators, and each unit resonator includes a transmission line including a first signal conducting portion, a second signal conducting portion, and a ground conducting portion corresponding to the first signal conducting portion and the second signal conducting portion, a first conductor that electrically connects the first signal conducting portion and the ground conducting portion, a second conductor that electrically connects the second signal conducting portion and the ground conducting portion, and at least one capacitor inserted between the first signal conducting portion and the second signal conducting portion in series with respect to a current flowing through the first signal conducting portion and the second signal conducting portion. 
     The transmission line, the first conductor, the second conductor may form a loop structure. 
     The at least two unit resonators may include a first unit resonator and a second unit resonator, the first unit resonator may be disposed on an upper plane and the second unit resonator is disposed on a lower plane, and the upper plane and the lower plane may be disposed at a predetermined distance away from each other. 
     An external circumference of the first unit resonator may be equal to an external circumference of the second unit resonator, and an area of an internal loop of the first unit resonator may be equal to an area of an internal loop of the second unit resonator. 
     A capacitor inserted into the first unit resonator may be disposed in an opposite direction to a direction in which a capacitor inserted into the second unit resonator is disposed. 
     The second unit resonator may be included in a loop of the first unit resonator. 
     The wireless power resonator may further include a matcher, disposed inside a loop formed by the transmission line, the first conductor, and the second conductor, to determine an impedance of the wireless power resonator. 
     In another general aspect, there is provided a wireless power resonator, including a transmission line including a first signal conducting portion, a second signal conducting portion, a ground conducting portion corresponding to the first signal conducting portion and the second signal conducting portion, a first conductor that electrically connects the first signal conducting portion and the ground conducting portion, a second conductor that electrically connects the second signal conducting portion and the ground conducting portion, and at least one capacitor inserted between the first signal conducting portion and the second signal conducting portion in series with respect to a current flowing through the first signal conducting portion and the second signal conducting portion, and the first signal conducting portion, the second signal conducting portion, the first conductor, and the second conductor form a plurality of turns. 
     The plurality of turns included in at least one transmission line may be disposed in the same plane. 
     The wireless power resonator may further include a matcher, disposed inside a loop formed by the transmission line, the first conductor, and the second conductor. 
     In still another general aspect, there is provided a wireless power resonator, including a first unit resonator and at least one second unit resonator having a size less than a size of the first unit resonator, and each resonance unit includes a transmission line including a first signal conducting portion, a second signal conducting portion, and a ground conducting portion corresponding to the first signal conducting portion and the second signal conducting portion, a first conductor that electrically connects the first signal conducting portion and the ground conducting portion, a second conductor that electrically connects the second signal conducting portion and the ground conducting portion, and at least one capacitor inserted between the first signal conducting portion and the second signal conducting portion in series with respect to a current flowing through the first signal conducting portion and the second signal conducting portion, and the at least one second unit resonator is disposed inside a loop of the first unit resonator. 
     The at least one second unit resonator may be disposed inside the loop of the first unit resonator at regular intervals. 
     Each unit resonator may further include a matcher that is disposed inside the loop formed by the transmission line, the first conductor, and the second conductor so as to determine an impedance of the wireless power resonator. 
     In yet another general aspect, there is provided a wireless power resonator, including at least two unit resonators forming magnetic fields in different directions, and each unit resonator includes a transmission line including a first signal conducting portion, a second signal conducting portion, and a ground conducting portion corresponding to the first signal conducting portion and the second signal conducting portion, a first conductor that electrically connects the first signal conducting portion and the ground conducting portion, a second conductor that electrically connects the second signal conducting portion and the ground conducting portion, and at least one capacitor inserted between the first signal conducting portion and the second signal conducting portion in series with respect to a current flowing through the first signal conducting portion and the second signal conducting portion. 
     The at least two unit resonators may be disposed to enable magnetic fields formed by the at least two unit resonators to be orthogonal to each other. 
     The current may flow through at least one resonator selected from among the at least two unit resonators. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a diagram illustrating an example of a wireless power transmission system including resonators for wireless power transmission. 
         FIG. 2  is a diagram illustrating an example of a resonator having a two-dimensional (2D) structure. 
         FIG. 3  is a diagram illustrating an example of a resonator having a three-dimensional (3D) structure. 
         FIG. 4  is a diagram illustrating an example of a resonator for wireless power transmission configured as a bulk type. 
         FIG. 5  is a diagram illustrating an example of a resonator for wireless power transmission configured as a hollow type. 
         FIG. 6  is a diagram illustrating an example of a resonator for wireless power transmission using a parallel-sheet. 
         FIG. 7  is a diagram illustrating an example of a resonator for wireless power transmission, including a distributed capacitor. 
         FIG. 8  are diagrams illustrating an example of a matcher used by a 2D resonator and an example of a matcher used by a 3D resonator. 
         FIG. 9  is a diagram illustrating an example of a first unit resonator and a second unit resonator included in a split-ring type resonator for wireless power transmission. 
         FIG. 10  is a diagram illustrating an example of a split-ring type resonator in 3D. 
         FIG. 11  is a diagram illustrating an example of a split-ring type resonator including two unit resonators having different sizes. 
         FIG. 12  is a diagram illustrating an example of a resonator for wireless power transmission, having a plurality of turns in a horizontal direction. 
         FIG. 13  is a diagram illustrating an example of a resonator for wireless power transmission, having a plurality of turns in a vertical direction. 
         FIG. 14  is a diagram illustrating an example of a resonator for wireless power transmission, including a relatively large unit resonator and relatively small unit resonators disposed inside a loop of the relatively large unit resonator. 
         FIG. 15  is a diagram illustrating an example of a 3D resonator for wireless power transmission, having an omni-directional characteristic. 
         FIG. 16  is a diagram illustrating an example of an equivalent circuit of a resonator for wireless power transmission of  FIG. 2 . 
         FIG. 17  is a diagram illustrating an example of an equivalent circuit of a composite right-left handed transmission line having a zeroth-order resonance characteristic. 
         FIG. 18  is a graph illustrating a zeroth-order resonance generated from a composite right-left handed transmission line. 
         FIG. 19  is a table illustrating an example of characteristics of a resonator for wireless power transmission. 
         FIGS. 20 through 22  are diagrams illustrating examples of a resonator for wireless power transmission. 
         FIG. 23  is a block diagram illustrating an example of a wireless power transmitter that is applicable to a source of  FIG. 1 . 
         FIG. 24  is a block diagram illustrating an example of a wireless power receiver that is applicable to a destination of  FIG. 1 . 
     
    
    
     BEST MODE FOR CARRYING OUT THE INVENTION 
     The following detailed description is provided to assist the reader in gaining a comprehensive understanding of the methods, apparatuses and/or systems described herein. Accordingly, various changes, modifications, and equivalents of the systems, apparatuses and/or methods described herein may be suggested to those of ordinary skill in the art. The progression of processing steps and/or operations described is an example; however, the sequence of and/or operations is not limited to that set forth herein and may be changed as is known in the art, with the exception of steps and/or operations necessarily occurring in a certain order. Also, descriptions of well-known functions and constructions may be omitted for increased clarity and conciseness. 
       FIG. 1  illustrates an example of a wireless power transmission system, including a resonator for wireless power transmission and a resonator for wireless power reception. 
     The wireless power transmission system using a resonance characteristic of  FIG. 1  may include a source  110  and a destination  120 . Here, the source  110  may wirelessly provide power to the destination  120  using a resonator configured as a helix coil structure or a resonator configured as a spiral coil structure. 
     Here, physical sizes of the resonator configured as the helix coil structure or the resonator configured as the spiral coil structure may be dependent upon a desired resonant frequency. For example, when the desired resonant frequency is 10 megahertz (Mhz), a diameter of the resonator configured as the helix coil structure may be determined to be about 0.6 meters (m), and a diameter of the resonator configured as the spiral coil structure may be determined to be about 0.6 m. In this example, as a desired resonant frequency decreases the diameter of the resonator configured as the helix coil structure and the diameter of the resonator configured as the spiral coil structure may need to be increased. 
     The change occurs in a physical size of a resonator due to a change in a resonant frequency is not exemplary. As an extreme example, when a resonant frequency is significantly low, a size of the resonator may be remarkably large, which may not be practical. When the resonant frequency is independent of the size of the resonant, the resonator may be exemplary. Also, a resonator that has a rational physical size and operates well irrespective of a high resonant frequency and a low resonant frequency may be an exemplary resonator. 
     Hereinafter, related terms will be described for concise understanding although the terms are well known. All the materials may have a unique magnetic permeability, that is, Mu, and a unique permittivity, that is, epsilon. The magnetic permeability indicates a ratio between a magnetic flux density occurring with respect to a given magnetic field in a corresponding material and a magnetic flux density occurring with respect to the given magnetic field in a vacuum state. The permittivity indicates a ratio between an electric flux density occurring with respect to a given electric field in a corresponding material and an electric flux density occurring with respect to the given electric field in a vacuum state. The magnetic permeability and the permittivity may determine a propagation constant of a corresponding material in a given frequency or a given wavelength. An electromagnetic characteristic of the corresponding material may be determined based on the magnetic permeability and the permittivity. In particular, a material having a magnetic permeability or a permittivity not found in nature and being artificially designed is referred to as a metamaterial. The metamaterial may be easily disposed in a resonance state even in a relatively large wavelength area or a relatively low frequency area. For example, even though a material size rarely varies, the metamaterial may be easily disposed in the resonance state. 
       FIG. 2  illustrates an example of a resonator having a 2D structure. 
     Referring to  FIG. 2 , the resonator having the 2D structure includes a transmission line, a capacitor  220 , a matcher  220 , and conductors  241  and  242 . The transmission line includes a first signal conducting portion  211 , a second signal conducting portion  212 , and a ground conducting portion  213 . 
     The capacitor  220  may be inserted in series between the first signal conducting portion  211  and the second signal conducting portion  212 , whereby an electric field may be confined within the capacitor  220 . Generally, the transmission line may include at least one conductor in an upper portion of the transmission line, and may also include at least one conductor in a lower portion of the transmission line. A current may flow through the at least one conductor disposed in the upper portion of the transmission line and the at least one conductor disposed in the lower portion of the transmission may be electrically grounded. Herein, a conductor disposed in an upper portion of the transmission line may be separated into and thereby be referred to as the first signal conducting portion  211  and the second signal conducting portion  212 . A conductor disposed in the lower portion of the transmission line may be referred to as the ground conducting portion  213 . 
     As shown in  FIG. 2 , the resonator  200  may have the 2D structure. The transmission line may include the first signal conducting portion  211  and the second signal conducting portion  212  in the upper portion of the transmission line, and may include the ground conducting portion  213  in the lower portion of the transmission line. The first signal conducting portion  211  and the second signal conducting portion  212  may be disposed to face the ground conducting portion  213 . The current may flow through the first signal conducting portion  211  and the second signal conducting portion  212 . 
     One end of the first signal conducting portion  211  may be shorted to a conductor  242 , and another end of the first signal conducting portion  211  may be connected to the capacitor  220 . One end of the second signal conducting portion  212  may be shorted to the conductor  241 , and another end of the second signal conducting portion  212  may be connected to the capacitor  220 . Accordingly, the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  may be connected to each other, whereby the resonator  200  may have an electrically closed-loop structure. The term “loop structure” may include a polygonal structure, for example, a circular structure, a rectangular structure, and the like. “Having a loop structure” may indicate being electrically closed. 
     The capacitor  220  may be inserted into an intermediate portion of the transmission line. Specifically, the capacitor  220  may be inserted into a space between the first signal conducting portion  211  and the second signal conducting portion  212 . The capacitor  220  may have a shape of a lumped element, a distributed element, and the like. In particular, a distributed capacitor having the shape of the distributed element may include zigzagged conductor lines and a dielectric material having a relatively high permittivity between the zigzagged conductor lines. 
     When the capacitor  220  is inserted into the transmission line, the resonator  200  may have a property of a metamaterial. The metamaterial indicates a material having a predetermined electrical property that cannot be discovered in nature and thus, may have an artificially designed structure. An electromagnetic characteristic of all the materials existing in nature may have a unique magnetic permeability or a unique permittivity. Most materials may have a positive magnetic permeability or a positive permittivity. In the case of most materials, a right hand rule may be applied to an electric field, a magnetic field, and a pointing vector and thus, the corresponding materials may be referred to as right handed materials (RHMs). However, the metamaterial has a magnetic permeability or a permittivity absent in nature and thus, may be classified into an epsilon negative (ENG) material, a mu negative (MNG) material, a double negative (DNG) material, a negative refractive index (NRI) material, a left-handed (LH) material, and the like, based on a sign of the corresponding permittivity or magnetic permeability. 
     When a capacitance of the capacitor inserted as the lumped element is appropriately determined, the resonator  200  may have the characteristic of the metamaterial. Since the resonator  200  may have a negative magnetic permeability by appropriately adjusting the capacitance of the capacitor  220 , the resonator  200  may also be referred to as an MNG resonator. Various criteria may be applied to determine the capacitance of the capacitor  220 . For example, the various criteria may include a criterion for enabling the resonator  200  to have the characteristic of the metamaterial, a criterion for enabling the resonator  200  to have a negative magnetic permeability in a target frequency, a criterion for enabling the resonator  200  to have a zeroth-order resonance characteristic in the target frequency, and the like. Based on at least one criterion among the aforementioned criteria, the capacitance of the capacitor  220  may be determined. 
     The resonator  200 , also referred to as the MNG resonator  200 , may have a zeroth-order resonance characteristic of having, as a resonance frequency, a frequency when a propagation constant is “0”. Since the resonator  200  may have the zeroth-order resonance characteristic, the resonance frequency may be independent with respect to a physical size of the MNG resonator  200 . By appropriately designing the capacitor  220 , the MNG resonator  200  may sufficiently change the resonance frequency. Accordingly, the physical size of the MNG resonator  200  may not be changed. 
     In a near field, the electric field may be concentrated on the capacitor  220  inserted into the transmission line. Accordingly, due to the capacitor  220 , the magnetic field may become dominant in the near field. The MNG resonator  200  may have a relatively high Q-factor using the capacitor  220  of the lumped element and thus, it is possible to enhance an efficiency of power transmission. Here, the Q-factor indicates a level of an ohmic loss or a ratio of a reactance with respect to a resistance in the wireless power transmission. It can be understood that the efficiency of the wireless power transmission may increase according to an increase in the Q-factor. 
     The MNG resonator  200  may include the matcher  220  for impedance-matching. The matcher  220  may appropriately adjust a strength of a magnetic field of the MNG resonator  200 . An impedance of the MNG resonator  200  may be determined by the matcher  220 . A current may flow in the MNG resonator  200  via a connector, or may flow out from the MNG resonator  200  via the connector. The connector may be connected to the ground conducting portion  213  or the matcher  220 . A physical connection may be formed between the connector and the ground conducting portion  213 , or between the connector and the matcher  220 . The power may be transferred through coupling without using a physical connection between the connector and the ground conducting portion  213  or the matcher  220 . 
     More specifically, as shown in  FIG. 2 , the matcher  220  may be positioned within the loop formed by the loop structure of the resonator  200 . The matcher  220  may adjust the impedance of the resonator  200  by changing the physical shape of the matcher  220 . For example, the matcher  220  includes a conductor  231  for the impedance-matching in a location separate from the ground conducting portion  213  by a distance h. The impedance of the resonator  200  may be changed by adjusting the distance h. 
     Although not illustrated in  FIG. 2 , a controller may be provided to control the matcher  220 . In this case, the matcher  220  may change the physical shape of the matcher  220  based on a control signal generated by the controller. For example, the distance h between a conductor  231  of the matcher  220  and the ground conducting portion  213  may increase or decrease based on the control signal. Accordingly, the physical shape of the matcher  220  may be changed whereby the impedance of the resonator  200  may be adjusted. The controller may generate the control signal based on varied factors, which will be described later. 
     As shown in  FIG. 2 , the matcher  220  may be configured as a passive element such as the conductor  231 . Depending on embodiments, the matcher  220  may be configured as an active element such as a diode, a transistor, and the like. When the active element is included in the matcher  220 , the active element may be driven based on the control signal generated by the controller, and the impedance of the resonator  200  may be adjusted based on the control signal. For example, a diode that is a type of the active element may be included in the matcher  220 . The impedance of the resonator  200  may be adjusted depending on whether the diode is in an on state or in an off state. 
     Although not illustrated in  FIG. 2 , a magnetic core may be further provided to pass through the MNG resonator  200 . The magnetic core may perform a function of increasing a power transmission distance. 
       FIG. 3  illustrates an example of a resonator having a 3D structure according to example embodiments. 
     Referring to  FIG. 3 , the resonator  200  having a 3D structure includes a transmission line and the capacitor  220 . The transmission line includes the first signal conducting portion  211 , the second signal conducting portion  212 , and the ground conducting portion  213 . The capacitor  220  may be inserted in series between the first signal conducting portion  211  and the second signal conducting portion  212  of the transmission line, whereby an electric field may be confined within the capacitor  220 . 
     As shown in  FIG. 3 , the resonator  200  may have the 3D structure. The transmission line includes the first signal conducting portion  211  and the second signal conducting portion  212  in an upper portion of the resonator  200 , and includes the ground conducting portion  213  in a lower portion of the resonator  200 . The first signal conducting portion  211  and the second signal conducting portion  212  may be disposed to face the ground conducting portion  213 . A current may flow in an x direction through the first signal conducting portion  211  and the second signal conducting portion  212 . Due to the current, a magnetic field H(W) may be formed in a −y direction. Alternatively, unlike the diagram of  FIG. 3 , the magnetic field H(W) may be formed in a +y direction. 
     One end of the first signal conducting portion  211  may be shorted to the conductor  242 , and another end of the first signal conducting portion  211  may be connected to the capacitor  220 . One end of the second signal conducting portion  212  may be shorted to the conductor  241 , and another end of the second signal conducting portion  212  may be connected to the capacitor  220 . Accordingly, the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  may be connected to each other, whereby the resonator  200  may have an electrically closed-loop structure. The term “loop structure” may include a polygonal structure, for example, a circular structure, a rectangular structure, and the like. “Having a loop structure” may indicate being electrically closed. 
     As shown in  FIG. 3 , the capacitor  220  may be inserted into a space between the first signal conducting portion  211  and the second signal conducting portion  212 . The capacitor  220  may have a shape of a lumped element, a distributed element, and the like. In particular, a distributed capacitor having the shape of the distributed element may include zigzagged conductor lines and a dielectric material having a relatively high permittivity between the zigzagged conductor lines. 
     As the capacitor  220  is inserted into the transmission line, the resonator  200  may have a property of a metamaterial. When a capacitance of the capacitor inserted as the lumped element is appropriately determined, the resonator  200  may have the characteristic of the metamaterial. Since the resonator  200  may have a negative magnetic permeability in a predetermined frequency band by appropriately adjusting the capacitance of the capacitor  220 , the resonator  200  may also be referred to as an MNG resonator. Various criteria may be applied to determine the capacitance of the capacitor  220 . For example, the various criteria may include a criterion for enabling the resonator  200  to have the characteristic of the metamaterial, a criterion for enabling the resonator  200  to have a negative magnetic permeability in a target frequency, a criterion for enabling the resonator  200  to have a zeroth-order resonance characteristic in the target frequency, and the like. Based on at least one criterion among the aforementioned criteria, the capacitance of the capacitor  220  may be determined. 
     The resonator  200 , also referred to as the MNG resonator  200 , may have a zeroth-order resonance characteristic of having, as a resonance frequency, a frequency when a propagation constant is “0”. Since the resonator  200  may have the zeroth-order resonance characteristic, the resonance frequency may be independent with respect to a physical size of the MNG resonator  200 . By appropriately designing the capacitor  220 , the MNG resonator  200  may sufficiently change the resonance frequency. Accordingly, the physical size of the MNG resonator  200  may not be changed. 
     Referring to the MNG resonator  200  of  FIG. 3 , in a near field, the electric field may be concentrated on the capacitor  220  inserted into the transmission line. Accordingly, due to the capacitor  220 , the magnetic field may become dominant in the near field. In particular, since the MNG resonator  200  having the zeroth-order resonance characteristic may have characteristics similar to a magnetic dipole, the magnetic field may become dominant in the near field. A relatively small amount of the electric field formed due to the insertion of the capacitor  220  may be concentrated on the capacitor  220  and thus, the magnetic field may become further dominant. The MNG resonator  200  may have a relatively high Q-factor using the capacitor  220  of the lumped element and thus, it is possible to enhance an efficiency of power transmission. 
     Also, the MNG resonator  200  includes a matcher  230  for impedance-matching. The matcher  230  may appropriately adjust the strength of magnetic field of the MNG resonator  200 . An impedance of the MNG resonator  200  may be determined by the matcher  230 . A current may flow in the MNG resonator  200  via a connector  240 , or may flow out from the MNG resonator  200  via the connector  240 . The connector  240  may be connected to the ground conducting portion  213  or the matcher  230 . 
     More specifically, as shown in  FIG. 3 , the matcher  230  may be positioned within the loop formed by the loop structure of the resonator  200 . The matcher  230  may adjust the impedance of the resonator  200  by changing the physical shape of the matcher  230 . For example, the matcher  230  includes the conductor  231  for the impedance-matching in a location separate from the ground conducting portion  213  by a distance h. The impedance of the resonator  200  may be changed by adjusting the distance h. 
     Although not illustrated in  FIG. 3 , a controller may be provided to control the matcher  230 . In this case, the matcher  230  may change the physical shape of the matcher  230  based on a control signal generated by the controller. For example, the distance h between the conductor  231  of the matcher  230  and the ground conducting portion  213  may increase or decrease based on the control signal. Accordingly, the physical shape of the matcher  230  may be changed whereby the impedance of the resonator  200  may be adjusted. The distance h between the conductor  231  of the matcher  230  and the ground conducting portion  213  may be adjusted using a variety of schemes. As one example, a plurality of conductors may be included in the matcher  230  and the distance h may be adjusted by adaptively activating one of the conductors. As another example, the distance h may be adjusted by adjusting the physical location of the conductor  231  up and down. The distance h may be controlled based on the control signal of the controller. The controller may generate the control signal using various factors. An example of the controller generating the control signal will be described later. 
     As shown in  FIG. 3 , the matcher  230  may be configured as a passive element such as the conductor  231 . Depending on embodiments, the matcher  230  may be configured as an active element such as a diode, a transistor, and the like. When the active element is included in the matcher  230 , the active element may be driven based on the control signal generated by the controller, and the impedance of the resonator  200  may be adjusted based on the control signal. For example, a diode that is a type of the active element may be included in the matcher  230 . The impedance of the resonator  200  may be adjusted depending on whether the diode is in an on state or in an off state. 
     Although not illustrated in  FIG. 3 , a magnetic core may be further provided to pass through the resonator  200  configured as the MNG resonator. The magnetic core may perform a function of increasing a power transmission distance. 
       FIG. 4  illustrates an example of a resonator for a wireless power transmission configured as a bulky type according to example embodiments. 
     Referring to  FIG. 4 , the first signal conducting portion  211  and the conductor  242  may be integrally formed instead of being separately manufactured and thereby be connected to each other. Similarly, the second signal conducting portion  212  and a conductor  241  may also be integrally manufactured. 
     When the second signal conducting portion  212  and the conductor  241  are separately manufactured and then are connected to each other, a loss of conduction may occur due to a seam  250 . Accordingly, the second signal conducting portion  212  and the conductor  241  may be connected to each other without using a separate seam, that is, may be seamlessly connected to each other. Accordingly, it is possible to decrease a conductor loss caused by the seam  250 . As another example, the second signal conducting portion  212  and the ground conducting portion  213  may be seamlessly and integrally manufactured. As another example, the first signal conducting portion  211  and the ground conducting portion  213  may be seamlessly and integrally manufactured. As another example, the first signal conducting portion  211  and the conductor  242  may be seamlessly manufactured. As another example, the conductor  242  and the ground conducting portion  213  may be seamlessly manufactured. 
     Referring to  FIG. 4 , a type of a seamless connection connecting at least two partitions into an integrated form is referred to as a bulky type. 
       FIG. 5  illustrates an example of a resonator for a wireless power transmission, configured as a hollow type according to example embodiments. 
     Referring to  FIG. 5 , each of the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and conductors  241  and  242  of the resonator  200  configured as the hollow type includes an empty space inside. 
     In a given resonance frequency, an active current may be modeled to flow in only a portion of the first signal conducting portion  211  instead of all of the first signal conducting portion  211 , the second signal conducting portion  212  instead of all of the second signal conducting portion  212 , the ground conducting portion  213  instead of all of the ground conducting portion  213 , and the conductors  241  and  242  instead of all of the conductors  241  and  242 . Specifically, when a depth of each of the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  is significantly deeper than a corresponding skin depth in the given resonance frequency, it may be ineffective. The significantly deeper depth may increase a weight or manufacturing costs of the resonator  200 . 
     Accordingly, in the given resonance frequency, the depth of each of the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  may be appropriately determined based on the corresponding skin depth of each of the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242 . When each of the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  has an appropriate depth deeper than a corresponding skin depth, the resonator  200  may become light, and manufacturing costs of the resonator  200  may also decrease. 
     For example, as shown in  FIG. 5 , the depth of the second signal conducting portion  212  may be determined as “d” mm and d may be determined according to 
             d   =       1       π   ⁢           ⁢   f   ⁢           ⁢   μσ         .           
Here, f denotes a frequency, μ denotes a magnetic permeability, and σ denotes a conductor constant. When the first signal conducting portion  211 , the second signal conducting portion  212 , the ground conducting portion  213 , and the conductors  241  and  242  are made of a copper and have a conductivity of 5.8×10 7  siemens per meter (S·m −1 ), the skin depth may be about 0.6 mm with respect to 10 kHz of the resonance frequency and the skin depth may be about 0.006 mm with respect to 100 MHz of the resonance frequency.
 
       FIG. 6  illustrates an example of a resonator for a wireless power transmission using a parallel-sheet according to example embodiments. 
     Referring to  FIG. 6 , the parallel-sheet may be applicable to each of the first signal conducting portion  211  and the second signal conducting portion  212  included in the resonator  200 . 
     Each of the first signal conducting portion  211  and the second signal conducting portion  212  may not be a perfect conductor and thus, may have a resistance. Due to the resistance, an ohmic loss may occur. The ohmic loss may decrease a Q-factor and also decrease a coupling effect. 
     By applying the parallel-sheet to each of the first signal conducting portion  211  and the second signal conducting portion  212 , it is possible to decrease the ohmic loss, and to increase the Q-factor and the coupling effect. Referring to a portion  270  indicated by a circle, when the parallel-sheet is applied, each of the first signal conducting portion  211  and the second signal conducting portion  212  includes a plurality of conductor lines. The plurality of conductor lines may be disposed in parallel, and may be shorted at an end portion of each of the first signal conducting portion  211  and the second signal conducting portion  212 . 
     As described above, when the parallel-sheet is applied to each of the first signal conducting portion  211  and the second signal conducting portion  212 , the plurality of conductor lines may be disposed in parallel. Accordingly, a sum of resistances having the conductor lines may decrease. Consequently, the resistance loss may decrease, and the Q-factor and the coupling effect may increase. 
       FIG. 7  illustrates an example of a resonator for a wireless power transmission, including a distributed capacitor according to example embodiments. 
     Referring to  FIG. 7 , the capacitor  220  included in the resonator for the wireless power transmission may be a distributed capacitor. A capacitor as a lumped element may have a relatively high equivalent series resistance (ESR). A variety of schemes have been proposed to decrease the ESR contained in the capacitor of the lumped element. According to an embodiment, by using the capacitor  220  as a distributed element, it is possible to decrease the ESR. As is known in the art, a loss caused by the ESR may decrease a Q-factor and a coupling effect. 
     As shown in  FIG. 7 , the capacitor  220  as the distributed element may have a zigzagged structure. For example, the capacitor  220  as the distributed element may be configured as a conductive line and a conductor having the zigzagged structure. 
     As shown in  FIG. 7 , by employing the capacitor  220  as the distributed element, it is possible to decrease the loss occurring due to the ESR. In addition, by disposing a plurality of capacitors as lumped elements, it is possible to decrease the loss occurring due to the ESR. Since a resistance of each of the capacitors as the lumped elements decreases through a parallel connection, active resistances of parallel-connected capacitors as the lumped elements may also decrease whereby the loss occurring due to the ESR may decrease. For example, by employing ten capacitors of 1 pF instead of using a single capacitor of 10 pF, it is possible to decrease the loss occurring due to the ESR. 
       FIG. 8  illustrates an example of a matcher used in a resonator provided in a 2D structure, an example of a matcher used in a resonator provided in a 3D structure. 
     Here, a diagram A in  FIG. 8  illustrates a portion of the 2D resonator of  FIG. 2  including the matcher, and a diagram B in  FIG. 8  illustrates a portion of the 3D resonator of  FIG. 3  including the matcher. 
     Referring to the diagram A in  FIG. 8 , the matcher includes the conductor  231 , a conductor  232 , and a conductor  233 . The conductors  232  and  233  may be connected to the ground conducting portion  213  and the conductor  231 . The impedance of the 2D resonator may be determined based on a distance h between the conductor  231  and the ground conducting portion  213 . The distance h between the conductor  231  and the ground conducting portion  213  may be controlled by the controller. The distance h between the conductor  231  and the ground conducting portion  213  may be adjusted using a variety of schemes. For example, the variety of schemes may include a scheme of adjusting the distance h by adaptively activating one of conductors, for example, the conductor  231 , a scheme of adjusting the physical location of the conductor  231  up and down, and the like. 
     Referring to the diagram B in  FIG. 8 , the matcher includes the conductor  231 , a conductor  232 , and a conductor  233 . The conductors  232  and  233  may be connected to the ground conducting portion  213  and the conductor  231 . The impedance of the 3D resonator may be determined based on a distance h between the conductor  231  and the ground conducting portion  213 . The distance h between the conductor  231  and the ground conducting portion  213  may be controlled by the controller. Similar to the matcher included in the 2D structured resonator, in the matcher  230  included in the 3D structured resonator, the distance h between the conductor  231  and the ground conducting portion  213  may be adjusted using a variety of schemes. For example, the variety of schemes may include a scheme of adjusting the distance h by adaptively activating one of conductors, for example, the conductor  231 , a scheme of adjusting the physical location of the conductor  231  up and down, and the like. 
     Although not illustrated in  FIG. 8 , the matcher may include an active element. A scheme of adjusting an impedance of a resonator using the active element may be similar as described above. For example, the impedance of the resonator may be adjusted by changing a path of a current flowing through the matcher using the active element. 
       FIG. 9  illustrates an example of a first unit resonator and a second unit resonator included in a split-ring type resonator for wireless power transmission. 
     Referring to  FIG. 9 , each of the first unit resonator  21  and the second unit resonator  22  included in the split-ring type resonator may include a transmission line and a capacitor inserted in series into a middle portion of the transmission line. The first unit resonator  21  may include a matcher having a thick of ‘e’ and a height of ‘c’. The second unit resonator  22  may additionally include a matcher, although not illustrated in  FIG. 9 . When the second unit resonator  22  additionally includes the matcher, the matcher of the first unit resonator  21  and the matcher of the second unit resonator  22  may be connected to each other through a ‘via(hole)’. In this example, when power is applied to one of the unit resonators  21  and  22 , the two unit resonators  21  and  22  may operate together. 
     In addition, the first unit resonator  21  has a height of ‘a’, and a width of ‘b’. The transmission line has a thickness of ‘d’. The matcher has a height of ‘c’ and a thickness of ‘e’. In this example, ‘a’ may be in a range from about 50 millimeters (mm) to 70 mm, ‘b’ may be in a range from about 30 mm to 50 mm, ‘c’ may be in a range from about 4 mm to 4.6 mm, ‘d’ may be in a range from about 4.5 mm to 5.5 mm, and ‘e’ may be in a range from about 1.7 mm to 2.3 mm. For example, ‘a’ may be 60 mm, ‘b’ may be 40 mm, ‘c’ may be 4.3 mm, ‘d’ may be 5 mm, and ‘e’ may be 2 mm. Here, ‘a’, ‘b’, ‘c’, ‘d’, and ‘e’ are merely examples of the size. That is, ‘a’ may be greater than 70 mm. A value of ‘h’ may be adaptively adjusted based on a desired resonant frequency, and the like. 
     The first unit resonator  21  and the second unit resonator  22  of  FIG. 9  may be stacked in two layers, and a resonator including the first unit resonator  21  and the second unit resonator  22  in two layers may be referred to as a split-ring type resonator. For example, the first unit resonator  21  may be disposed on one plane, and the second unit resonator  22  may be disposed on another plane located at a predetermined distance away from the plane. 
     When an external circumference of the first unit resonator  21  of  FIG. 9  is equal to an external circumference of the second unit resonator  22 , and an area of an internal loop of the first unit resonator  21  is equal to an area of an internal loop of the second unit resonator  22 , a mutual coupling between the first unit resonator  21  and the second unit resonator  22  may be maximized. 
     The capacitor of the first unit resonator  21  and the capacitor of the second unit resonator  22  may be inserted in the same direction, or in opposite directions. A case where the capacitor of the first unit resonator  21  is inserted in an opposite direction to the capacitor of the second unit resonator  22  will be described with reference to  FIG. 10 . 
       FIG. 10  illustrates an example of a split-ring type resonator in three-dimensions. 
     Referring to  FIG. 10 , the second unit resonator is disposed below the first unit resonator  21 , and the split-ring type resonator may include the first unit resonator  21  and the second unit resonator  22  stacked in two layers. 
     As shown in  FIG. 10 , a capacitor of the first unit resonator  21  and a capacitor of the second unit resonator  22  may be inserted in opposite directions. Although not illustrated in  FIG. 10 , the capacitor of the first unit resonator  21  and the capacitor of the second unit resonator  22  may be inserted in the same direction. 
     As illustrated in  FIG. 10 , when an external circumference of the first unit resonator  21  is equal to an external circumference of the second unit resonator  22 , and an area of an internal loop of the first unit resonator  21  is equal to an area of an internal loop of the second unit resonator  22 , a mutual coupling between the first unit resonator  21  and the second unit resonator  22  may be maximized. 
       FIG. 11  illustrates an example of a split-ring type resonator including two unit resonators having different sizes. 
     Referring to  FIG. 11 , the split-ring type resonator may include the first unit resonator  21  and the second unit resonator  22 , having different sizes from one another. That is, in the split-ring type resonator, the second unit resonator  22  may be included inside a loop of the first unit resonator  21 . In this example, the first unit resonator  21  and the second unit resonator  22  may be disposed on the same plane or on different planes from one another. 
     In  FIG. 11 , C 1  denotes a capacitor inserted into the first unit resonator  21  and C 1  denotes a capacitor inserted into the second unit resonator  22 . 
       FIG. 12  illustrates an example of a resonator for wireless power transmission, having a plurality of turns in the horizontal direction. 
     Referring to  FIG. 12 , the resonator having the plurality of turns in the horizontal direction may include the plurality of turns formed in the horizontal direction. In particular, referring to a resonator including 2 turns in  FIG. 12 , conductors in an upper portion of the resonator may correspond to a first signal conducting portion and a second signal conducting portion included in a transmission line, and conductors in a lower portion of the resonator may correspond to a ground conducting portion included in the transmission line. Conductors in the left portion of the resonator may correspond to first conductors connecting the first signal conducting portion and the ground conducting portion, and conductors in the right portion of the resonator may correspond to second conductors connecting the second signal conducting portion and the ground conducting portion. In this example, the first signal conducting portion, the second signal conducting portion, the first conductor, and the second conductor may include the 2 turns. In this manner, referring to a resonator including 3 turns in  FIG. 12 , a first signal conducting portion, a second signal conducting portion, a first conductor, and a second conductor included in the resonator may include the 3 turns. Here, sizes of turns may be the same or may be different from each other. 
     Since the plurality of turns is formed on substantially the same plane, the turns may be regarded to be formed in the horizontal direction. An inductance value of the resonator may increase due to the plurality of turns formed in the horizontal direction. Accordingly, a capacitance of a capacitor with respect to the inductance may be relatively decreased. Therefore, an effect of an ESR may be decreased, and the resonator for wireless power transmission, having the plurality of turns, may be applicable to a mobile device, for example, a portable phone. 
       FIG. 13  illustrates an example of a resonator for wireless power transmission, having a plurality of turns in the vertical direction. 
     Referring to a front view of the resonator in  FIG. 13 , the resonator having the plurality of turns in the vertical direction may include the resonator of  FIG. 2 . Referring to a side view of the resonator in  FIG. 13 , the plurality of turns is formed in the vertical direction. Here, the plurality of turns may be electrically connected. 
     Referring to  FIG. 13 , the plurality of turns may exist in different planes, respectively. In this example, sizes of the turns may be the same or may be different from each other. Since the plurality of turns exists in different planes, the plurality of turns may be regarded to be formed in the vertical direction. The resonator having the plurality of turns may be in a 3-layered structure. Due to the 3-layered structure, inductance with respect to a given sectional area may increase, and a capacitance of a capacitor may be relatively reduced. Accordingly, the resonator may be less affected by an ESR, and well may be applicable to a mobile device, for example, a portable phone. In addition, the resonator of  FIG. 13  may minimize an effect to an ambient environment, when compared to the resonator of  FIG. 12 , having the plurality of turns in the horizontal direction. 
     Based on the concept of a parallel-sheet, the plurality of turns in  FIG. 12  and  FIG. 13  may be shorted to a single ground plane. In this example, when the plurality of turns illustrated in  FIGS. 12 and 13  are shorted to one ground plane, a resistance loss may be minimized and thus, a Q factor may be improved. 
       FIG. 14  illustrates an example of a resonator for wireless power transmission, including a relatively large unit resonator and relatively small unit resonators disposed inside a loop of the relatively large unit resonator. 
     Referring to  FIG. 14 , the resonator for wireless power transmission may include a first unit resonator having a width of ‘a’ and a height of b′, and six second unit resonators which are smaller than the first unit resonator. Here, a number of second unit resonators may be variable. The six second unit resonators may be located inside a loop of the first unit resonator, and may be disposed inside the loop of the first unit resonator at regular intervals. Here, ‘a’ may be about 260 mm, ‘b’ may be about 150 mm, ‘c’ may be about 27.5 mm, and ‘d’ may be about 16.7 mm. 
     The resonator of  FIG. 14  may have a high effective magnetic permeability (mu), and may increase a power transmission gain. The second unit resonator may perform a function of effectively increasing the mu of the resonator and thus, the overall value of mu for the resonator may increase. Referring to the resonator of  FIG. 14 , mu may be improved based on an alignment of the first unit resonator and the second unit resonators, as opposed to based on a material. 
       FIG. 15  illustrates an example of a 3D resonator for wireless power transmission, having an omni-directional characteristic. 
     Referring to resonator  1  of  FIG. 15 , a magnetic field by a first unit resonator and a magnetic field by a second unit resonator may be generated in different directions. In particular, the magnetic field by the first unit resonator and the magnetic field by the second unit resonator may be orthogonal to each other and thus, the magnetic fields may not be coupled to one another. Accordingly, resonator  1  of  FIG. 15  may perform omni-directional power transmission. 
     Referring to resonator  2  of  FIG. 15 , six surfaces included in a regular hexahedron may correspond to unit resonators, respectively. Resonator  2  of  FIG. 15  may perform omni-directional power transmission, and may adaptively adjust a strength of the magnetic field. 
     A resonator according to example embodiments may have a structure of resonator  3  of  FIG. 15 , and the resonator may perform omni-directional power transmission. 
     The resonator according to example embodiments may have a spheral structure, for example, resonator  4  of  FIG. 15 . Referring to resonator  4  of  FIG. 15 , an external side of the sphere may be enclosed by transmission lines, a cross feeder may be disposed inside the sphere. The cross feeder may include feeders, and power may be transmitted in a direction corresponding to an activated feeder among the feeders. In particular, when all the feeders are activated, power may be transmitted in omni-directions. 
       FIG. 16  illustrates an example of an equivalent circuit of a resonator for wireless power transmission of  FIG. 2 . 
     The resonator for wireless power transmission in  FIG. 2  may be modeled to be an equivalent circuit of  FIG. 16 . In the equivalent circuit of  FIG. 16 , CL denotes a capacitor inserted, in a form of lumped elements, into a middle portion of the transmission line of  FIG. 2 . 
     In this example, the resonator for wireless power transmission in  FIG. 2  may have a zeroth-order resonance characteristic. That is, when a propagation constant is ‘0’, the resonator for wireless power transmission may have a resonant frequency of ω MZR . In this example, ω MZR  may be expressed by Equation 1. Here, MZR may denote a Mu zero resonator. 
     
       
         
           
             
               
                 
                   
                     ω 
                     MZR 
                   
                   = 
                   
                     1 
                     
                       
                         
                           L 
                           R 
                         
                         ⁢ 
                         
                           C 
                           L 
                         
                       
                     
                   
                 
               
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     1 
                   
                   ] 
                 
               
             
           
         
       
     
     Referring to Equation 1, ω MZR  that is, the resonant frequency of the resonator, may be determined based on 
                 L   R     /     C   L       ,         
and may ω MZR  be independent of a physical size of the resonator. Accordingly, the physical size of the resonator is independent of ω MZR  and thus, the physical size of the resonator may be sufficiently reduced.
 
       FIG. 17  illustrates an example of an equivalent circuit of a composite right-left handed transmission line having a zeroth-order resonance characteristic. Although the resonator for wireless power transmission is based on an MNG transmission line, the MNG transmission line will be described using the composite right-left handed transmission line. 
     Referring to  FIG. 17 , an equivalent circuit of the composite right-left handed transmission line may include a basic transmission line, and may additionally include 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412  that is a series-capacitor, and
 
                 L   L   ′     /   Δ     ⁢           ⁢   z           422  that is shunt-inductor. Here,
 
                 L   R   ′     /   Δ     ⁢           ⁢   z           411  and
 
                 C   R   ′     /   Δ     ⁢           ⁢   z           421  may denote an inductor component and a capacitor component of the basic transmission line, respectively.
 
     In this example, impedance Z′  410  may be a sum of a component corresponding to 
                 L   R   ′     /   Δ     ⁢           ⁢   z           411  and a component corresponding to
 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412 , and admittance Y′  420  may be a sum of a component corresponding to
 
                 C   R   ′     /   Δ     ⁢           ⁢   z           421  and a component corresponding to
 
                 L   L   ′     /   Δ     ⁢           ⁢   z           422 .
 
     Accordingly, impedance Z′  410  and admittance Y  420  may be expressed by Equation 2. 
     
       
         
           
             
               
                 
                   
                     
                       Z 
                       ′ 
                     
                     = 
                     
                       j 
                       ⁡ 
                       
                         ( 
                         
                           
                             ω 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             
                               L 
                               R 
                               ′ 
                             
                           
                           - 
                           
                             1 
                             
                               ω 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 C 
                                 L 
                                 ′ 
                               
                             
                           
                         
                         ) 
                       
                     
                   
                   ⁢ 
                   
                     
 
                   
                   ⁢ 
                   
                     
                       Y 
                       ′ 
                     
                     = 
                     
                       j 
                       ⁡ 
                       
                         ( 
                         
                           
                             ω 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             
                               C 
                               R 
                               ′ 
                             
                           
                           - 
                           
                             1 
                             
                               ω 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 L 
                                 L 
                                 ′ 
                               
                             
                           
                         
                         ) 
                       
                     
                   
                 
               
               
                 
                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     2 
                   
                   ] 
                 
               
             
           
         
       
     
     Referring to Equation 2, a resonant frequency, at which an amplitude of impedance Z  410  or admittance Y′  420  is minimized, may be adjusted by appropriately adding 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412  and
 
                 L   L   ′     /   Δ     ⁢           ⁢   z           422  to the transmission line. Also, the composite right-left handed transmission line has a zeroth-order resonance characteristic. That is, a resonant frequency of the composite right-left handed transmission line may be a frequency when a propagation constant is ‘0’.
 
     When only 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412  is added to the basic transmission line, the transmission line may have a negative value of mu in a predetermined frequency band and thus, may be referred to as an MNG transmission line.
 
     Also, when only 
                 L   L   ′     /   Δ     ⁢           ⁢   z           422  is added to the basic transmission line, the transmission line may have a negative permeability and thus, may be referred to as an ENG transmission line. The MNG transmission line and the ENG transmission line may also have a zeroth-order resonance characteristic.
 
     A MNG resonator according to example embodiments may include 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412  so that a magnetic field is dominant in a near field. That is, an electric field is concentrated on
 
                 C   L   ′     /   Δ     ⁢           ⁢   z           412  in the near field and thus, the magnetic field may be dominant in the near field.
 
     Also, the MNG resonator may have a zeroth-order resonance characteristic in the same manner as the composite right-left handed transmission line and thus, the MNG resonator may be manufactured to be small, irrespective of the resonant frequency. 
       FIG. 18  is a graph illustrating a zeroth-order resonance generated from a composite right-left handed transmission line. 
     Referring to  FIG. 18 , the composite right-left handed transmission line may have a resonant frequency of A and a resonant frequency of B. In this example, a propagation constant (β) corresponding to A and B is ‘0’ and thus, the composite right-left handed transmission line may have the zeroth-order resonance characteristic. 
     Similar to the composite right-left handed transmission line, an MNG transmission line and an ENG transmission line may also have the zeroth-order resonance characteristic. For example, a resonant frequency of the MNG transmission line may be A, and a resonant frequency of the ENG transmission line may be B. Accordingly, the MNG resonator may be manufactured to have a sufficiently small size. 
       FIG. 19  is a table illustrating characteristics of a resonator for wireless power transmission. 
     Referring to  FIG. 19 , a magnetic field is more dominant than an electric field in a near field of an MNG resonator. Also, the MNG resonator may improve a power transmission efficiency through a magnetic field coupling. 
     The MNG resonator may be manufactured in a 3D structure, and may aim for a high Q factor. The MNG resonator may be used for wireless power transmission in a short distance. 
       FIGS. 20 through 22  illustrate examples of a resonator for wireless power transmission. 
     Referring to  FIG. 20 , the resonator for wireless power transmission may include a plurality of transmission lines  710 ,  720 , and  730  which are connected in series. In this example, a plurality of capacitors  711 ,  721 ,  731  may be inserted into the plurality of transmission lines  710 ,  720 , and  730 , respectively. 
     Referring to  FIG. 21 , the resonator for wireless power transmission may have a spiral structure. That is, a plurality of transmission lines may be connected to each other so as to be configured as the spiral structure, and a plurality of capacitors may be inserted into the plurality of transmission lines, respectively. 
     Referring to  FIG. 22 , the resonator for wireless power transmission may include a plurality of transmission lines  910 ,  920 , and  930  which are connected to each other in parallel. 
     In addition to examples of  FIG. 20 through 22 , the resonator may be manufactured in various shapes. 
       FIG. 23  illustrates a configuration of a wireless power transmitter that is applicable to a source of  FIG. 1 . 
     Referring to  FIG. 23 , a wireless power transmitter  1000  may include a resonator  1010  and a pre-processor  1020 . 
     A wireless power transmission resonator  1010  may be a resonator described with respect to  FIGS. 1 through 22 , and power may be wirelessly transmitted using a wave propagated by the wireless power transmission resonator  1010 . 
     The pre-processor  1020  may generate a current and a frequency for wireless power transmission, using energy supplied from a power supplier existing inside or outside the wireless power transmitter  1000 . 
     In particular, the pre-processor  1020  may include an alternating current/direct current (AC/DC) converter  1021 , a frequency generator  1022 , a power amplifier  1023 , a controller  1024 , and a detector  1025 . 
     The AC/DC converter  1021  may convert AC energy supplied from the power supplier into DC energy or a DC current. In this example, the frequency generator  1022  may generate a desire frequency, that is, a desired resonant frequency, based on the DC energy or the DC current, and may generate a current having the desired frequency. The current having the desired frequency may be amplified by the power amplifier  1023 . 
     The controller  1024  may generate a control signal to control an impedance of the wireless power transmission resonator  1010 , and may adjust a frequency generated by the frequency generator  1022 . For example, an optimal frequency, at which a power transmission gain, a coupling efficiency, and the like are maximized, may be selected from among frequency bands. 
     The detector  1025  may detect a distance between the wireless power transmission resonator  1010  and a wireless power reception resonator of a wireless power receiver, a reflection coefficient of a wave radiated from the wireless power transmission resonator  1010  to the wireless power reception resonator, a power transmission gain between the wireless power transmission resonator  1010  and the wireless power reception resonator, a coupling efficiency between the wireless power transmission resonator  1010  and the wireless power reception resonator, or the like. 
     In this example, the controller  1024  may generate a control signal that adjusts an impedance of the wireless power transmission resonator  1010  based on the distance, the reflection coefficient, the power transmission gain, the coupling efficiency, and the like, or that controls a frequency generated by the frequency generator  1022 . 
       FIG. 24  illustrates a configuration of a wireless power receiver that is applicable to a destination of  FIG. 1 . 
     Referring to  FIG. 24 , a wireless power receiver  1100  may include a wireless power reception resonator  1110 , a rectifier  1120 , a detector  1130 , and a controller  1140 . 
     The wireless power reception resonator  1110  may be a resonator described with reference to  FIGS. 1 through 22 , and may receive a wave propagated by a wireless power transmitter. 
     The rectifier  1120  may convert power received by the wave into DC energy, and all or a portion of the DC energy may be provided to a target device. 
     The detector  1130  may detect a distance between the wireless power transmission resonator and the wireless power reception resonator  1110  of the wireless power receiver  1100 , a reflection coefficient of a wave radiated from the wireless power transmission resonator to the wireless power reception resonator  1100 , a power transmission gain between the wireless power transmission resonator and the wireless power reception resonator  1100 , a coupling efficiency between the wireless power transmission resonator and the wireless power reception resonator  1100 , or the like. 
     The controller  1140  may generate a control signal to control an impedance of the wireless power reception resonator  1100  based on the distance between the wireless power transmission resonator and the wireless power reception resonator  1110  of the wireless power receiver  1100 , the reflection coefficient of a wave radiated from the wireless power transmission resonator to the wireless power reception resonator  1100 , the power transmission gain between the wireless power transmission resonator and the wireless power reception resonator  1100 , the coupling efficiency between the wireless power transmission resonator and the wireless power reception resonator  1100 , or the like. 
     Although a few embodiments of the present invention have been shown and described, the present invention is not limited to the described embodiments. Instead, it would be appreciated by those skilled in the art that changes may be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the claims and their equivalents.