Patent Publication Number: US-RE37490-E

Title: Electronic caliper using a reduced offset induced current position transducer

Description:
BACKGROUND OF THE INVENTION 
     1. Field of Invention 
     This invention relates to an electronic caliper. More particularly this invention is directed to electronic calipers using a reduced offset induced current position transducer. 
     2. Description of Related Art 
     U.S. patent application Ser. No. 08/645,483 filed May 13, 1996, and incorporated herein in its entirety, discloses an electronic caliper using an inductive position transducer. 
     The operation of the electronic caliper using the inductive position transducer described in the application Ser. No. &#39;483 is generally shown in FIGS. 1,  2 , and  3 . As shown in FIG. 1, an inductive caliper  100  includes an elongated beam  102 . The elongated beam  102  is a rigid or semi-rigid bar having a generally rectangular cross section. A groove  106  is formed in an upper surface of the elongated beam  102 . An elongated measuring scale  104  is rigidly bonded to the elongated beam  102  in the groove  106 . The groove  106  is formed in the beam  102  at a depth about equal to the thickness of the scale  104 . Thus, the top surface of the scale  104  is very nearly coplanar with the top edges of beam  102 . 
     A pair of laterally projecting, fixed jaws  108  and  110  are integrally formed near a first end  112  of the beam  102 . A corresponding pair of laterally projecting movable jaws  116  and  118  are formed on a slider assembly  120 . The outside dimensions of an object are measured by placing the object between a pair of engagement surfaces  114  on the jaws  108  and  116 . Similarly, the inside dimensions of an object are measured by placing the jaws  110  and  118  within an object. The engagement surfaces  122  of the jaws  110  and  118  are positioned to contact the surfaces on the object to be measured. 
     The engagement surfaces  122  and  114  are positioned so that when the engagement surfaces  114  of the jaws  108  and  116  are contacting each other, the engagement surfaces  122  of the jaws  110  and  118  are aligned with each other. In this position, the zero position (not shown), both the outside and inside dimensions measured by the caliper  100  should be zero. 
     The caliper  100  also includes a depth bar  124  which is attached to the slider assembly  120 . The depth bar  124  projects longitudinally from the beam  102  and terminates at an engagement end  126 . The length of the depth bar  124  is such that the engagement end  126  is flush with a second end  128  of the beam  102  when the caliper  100  is at the zero position. By resting the second end  128  of the beam  102  on a surface in which a hole is formed and extending the depth bar  124  into the hole until the end  126  touches the bottom of the hole, the caliper  100  is able to measure the depth of the hole. 
     Whether a measurement is made using the outside measuring jaws  108  and  116 , the inside measuring jaws  110  and  118 , or the depth bar  124 , the measured dimension is displayed on a conventional digital display  134 , which is mounted in a cover  136  of the caliper  100 . A pair of push button switches  130  and  132  are also mounted in the cover  136 . The switch  130  turns on and off a signal processing and display electronic circuit  160  of the slider assembly  120 . The switch  132  is used to reset the display  134  to zero. 
     As shown in FIG. 1, the slider assembly  120  includes a base  138  with a guiding edge  140 . The guiding edge  140  contacts a side edge  146  of the elongated beam  102  when the slider assembly  120  straddles the elongated beam  102 . This ensures accurate operation of the caliper  100 . A pair of screws  144  bias a resilient pressure bar  146  against a mating edge of the beam  102  to eliminate free play between the slider assembly  120  and the elongated beam  102 . 
     The depth bar  124  is inserted into a depth bar groove  148  formed on an underside of the elongated beam  102 . The depth bar groove  148  extends along the underside of the elongated beam  102  to provide clearance for the depth bar  124 . The depth bar  124  is held in the depth bar groove  148  by an end stop  150 . The end stop  150  is attached to the underside of the beam  102  at the second end  128 . The end stop  150  also prevents the slider assembly  120  from inadvertently disengaging from the elongated beam  102  at the second end  128  during operation. 
     The slider assembly  120  also includes a read head assembly  152  mounted on the base  138  above the elongated beam  102 . Thus, the base  138  and read head assembly  152  move as a unit. The read head assembly  152  includes a substrate  154  such as a conventional printed circuit board. The substrate  154  bears an inductive read head  158  on its lower surface. A signal processing and display electronic circuit  160  is mounted on an upper surface of the substrate  154 . A resilient seal  156  is compressed between the cover  136  and the substrate  154  to prevent contamination of the signal processing and display electronic circuit  160 . 
     As shown in FIG. 2, the read head  158  is covered by a thin, durable, insulative coating  162 , which is preferably approximately 50 microns thick. 
     The scale  104  is preferably an elongated printed circuit board (PCB)  164 . As shown in FIG. 1, a set of magnetic flux modulators  166  are spaced apart along the PCB  164  in a periodic pattern. The flux modulators  166  are preferably formed of copper. The flux modulators  166  are preferably formed according to conventional printed circuit board manufacturing techniques, although many other methods of fabrication may be used. As shown in FIG. 2, a protective insulating layer  168  (preferably being at most 100 microns thick) covers the flux modulators  166 . The protective layer  168  can include printed markings, as shown in FIG.  1 . 
     The slider assembly  120  carries the read head  158  so that it is slightly separated from the beam  102  by an air gap  170  formed between the insulative coatings  162  and  168 . The air gap  170  is preferably on the order of 0.5 mm. Together, the read head  158  and the flux modulators  166  form an inductive transducer. 
     As shown in FIG. 3, the magnetic flux modulators  166  are distributed along a measuring axis  174  of the elongated beam  102  at a pitch equal to a wavelength λ, which is described in more detail below. The flux modulators  166  have a nominal width along the measuring axis  174  of λ/2. The flux modulators  166  have a width d in a direction perpendicular to the measuring axis  174 . 
     The read head  158  includes a generally square transmitter winding  176  that is connected to a drive signal generator  178 . The drive signal generator  178  provides a time varying drive signal to the transmitter winding  176 . The time varying drive signal preferably results in a sinusoidal signal in the transmitter winding  176 , and more preferably an exponentially decaying sinusoidal signal. When the time varying drive signal is applied to the transmitter winding  176 , the time varying current flowing in the transmitter winding  176  generates a time varying, or changing, magnetic field. Because the transmitter winding  176  is generally rectangularly shaped, the generated magnetic field is generally constant within a flux region inside the transmitter winding  176 . 
     The read head  158  further includes a first receiver winding  180  and a second receiver winding  182  positioned on the read head  158  within the flux region inside the transmitter winding  176 . Each of the first receiver winding  180  and the second receiver winding  182  is formed by a plurality of first loop segments  184  and second loop segments  186 . The first loop segments  184  are formed on a first surface of a layer of the printed circuit board  154 . The second loop segments  186  are formed on another surface of the layer of the printed circuit board  154 . The layer of the printed circuit board  154  acts as an electrical insulation layer between the first loop segments  184  and the second loop segments  186 . Each end of the first loop segments  184  is connected to one end of one of the second loop segments  186  through feed-throughs  188  formed in the layer of the printed circuit board  154 . 
     The first and second loop segments  184  and  186  are preferably sinusoidally shaped. Accordingly, as shown in FIG. 3 the first and second loop segments  184  and  186  forming each of the receiver windings  180  and  182  form a sinusoidally shaped periodic pattern having a wavelength λ. Each of the receiver windings  180  and  182  are thus formed having a plurality of loops  190  and  192 . 
     The loops  190  and  192  in each of the first and second receiver windings  180  and  182  have a width along the measuring axis  174  equal to λ/2. Thus, each pair of adjacent loops  190  and  192  has a width equal to λ. Furthermore, the first and second loop segments  184  and  186  go through a full sinusoidal cycle in each pair of adjacent loops  190  and  192 . Thus, λ corresponds to the sinusoidal wavelength of the first and second receiver windings  180  and  182 . Furthermore, the second receiver winding  182  is offset by λ/4 from the first receiver winding  180  along the measuring axis  174 . That is, the first and second receiver windings  180  and  182  are in quadrature. 
     The changing drive signal from the drive signal generator  178  is applied to the transmitter winding  176  such that current flows in a transmitter winding  176  from a first terminal  176   a,  through the transmitter winding  176  and out through a second terminal  176   b . Thus, the magnetic field generated by the transmitter winding  176  descends into the plane of FIG. 3 within the transmitter winding  176  and rises up out of the plane of FIG. 3 outside the transmitter winding  176 . Accordingly, the changing magnetic field within the transmitter winding  176  generates an induced electromagnetic force (EMF) in each of the loops  190  and  192  formed in the receiver windings  180  and  182 . 
     The loops  190  and  192  have opposite winding directions. Thus, the EMF induced in the loops  190  has a polarity that is opposite to the polarity of the EMF induced in the loops  192 . The loops  190  and  192  enclose the same area and thus nominally the same amount of magnetic flux. Therefore, the absolute magnitude of the EMF generated in each of the loops  190  and  192  is nominally the same. 
     There are preferably equal numbers of loops  190  and  192  in each of the first and second receiver windings  180  and  182 . Thus, the positive polarity EMF induced in the loops  190  is exactly offset by the negative polarity EMF induced in the loops  192 . Accordingly, the net nominal EMF on each of the first and second receiver windings  180  and  182  is zero. Thus, no signal should be output from the first and second receiver windings  180  and  182  as a result solely of the direct coupling from the transmitter winding  176  to the receiver windings  180  and  182 . 
     When the read head  158  is placed in proximity to the PCB  164 , the changing magnetic flux generated by the transmitter winding  176  also passes through the flux modulators  166 . The flux modulators  166  modulate the changing magnetic flux and can be either flux enhancers or flux disrupters. 
     When the flux modulators  166  are provided as flux disrupters, the flux modulators  166  are formed as conductive plates or thin conductive films on the PCB  164 . As the changing magnetic flux passes through the conductive plates or thin films, eddy currents are generated in the conductive plates or thin films. These eddy currents in turn generate magnetic fields having a field direction that is opposite to that of the magnetic field generated by the transmitter winding  176 . Thus, in areas proximate to each of the flux disrupter-type flux modulators  166 , the net magnetic flux is less than the net magnetic flux in areas distant from the flux disrupter type flux modulators  166 . 
     When the scale PCB  164  is positioned relative to the read head  158  such that the flux disrupters  166  are aligned with the positive polarity loops  190  of the receiver winding  180 , the net EMF generated in the positive polarity loops  190  is reduced compared to the net EMF generated in the negative polarity loops  192 . Thus, the receiver winding  190  becomes unbalanced and has a net negative signal across its output terminals  180   a  and  180   b.    
     Similarly, when the flux disrupters  166  are aligned with the negative polarity loops  192 , the net magnetic flux through the negative polarity loops  192  is disrupted or reduced. Thus, the net EMF generated in the negative polarity loops  192  is reduced relative to the net EMF generated in the positive polarity loops  190 . Thus, the first receiver winding  180  has a net positive signal across its output terminals  180   a  and  180   b.    
     When the flux modulators  166  are provided as flux enhancers, this result is exactly reversed. The flux enhancer type flux modulators  166  are formed by portions of high magnetic permeability material provided in or on the scale member  104 , in place of the conductive plates of PCB  164 . The magnetic flux generated by the transmitter winding  176  preferentially passes through the high magnetic permeability flux enhancer type flux modulators  166 . That is, the density of the magnetic flux within the flux enhancers  166  is enhanced, while the flux density in areas outside the flux enhancers  166  is reduced. 
     Thus, when the flux enhancers  166  are aligned with the positive polarity loops  190  of the second receiver winding  182 , the flux density through the positive polarity loops  190  is greater than a flux density passing through the negative polarity loops  192 . Thus, the net EMF generated in the positive polarity  190  increases, while the net EMF induced in the negative polarity loops  192  decreases. This appears as a positive signal across the terminals  182   a  and  182   b  of the second receiver winding  182 . 
     When the flux enhancers  166  are aligned with the negative polarity loops  192 , the negative polarity loops  192  generate an enhanced EMF relative to the EMF induced in the positive polarity loops  190 . Thus, a negative signal appears across the terminals  182   a  and  182   b  of the second receiver winding  180 . It should also be appreciated that, as outlined in the incorporated reference, both the flux enhancing and flux disrupting effects can be combined in a single scale, where the flux enhancers and flux disrupters are interleaved along the length of the scale  104 . This would act to enhance the modulation of the induced EMF, because the effects of both types of flux modulators additively combine. 
     As indicated above, the width and height of the flux modulators  166  are nominally λ/2 and d, respectively, while the pitch of the flux modulators  166  is nominally λ. Similarly, the wavelength of the periodic pattern formed in the first and second receiver windings  180  and  182  is nominally λ and the height of the loops  190  and  192  is nominally d. Furthermore, each of the loops  190  and  192  enclose a nominally constant area. 
     FIG. 4A shows the position-dependent output from the positive polarity loops  190  as the flux modulators  166  move relative to the positive polarity loops  190 . Assuming the flux modulators  166  are flux disrupters, the minimum signal amplitude corresponds to those positions where the flux disrupters  166  exactly align with the positive polarity loops  190 , while the maximum amplitude positions correspond to the flux disrupters  166  being aligned with the negative polarity loops  192 . 
     FIG. 4B shows the signal output from each of the negative polarity loops  192 . As with the signal shown in FIG. 4A, the minimum signal amplitude corresponds to those positions where the flux disrupters  166  exactly align with the positive polarity loops  190 , while the maximum signal output corresponds to those positions where the flux disrupters exactly align with the negative polarity loops  192 . It should be appreciated that if flux enhancers were used in place of flux disrupters, the minimum signal amplitudes in FIGS. 4A and 4B would correspond to the flux enhancers  166  aligning with the negative polarity loops  192 , while the maximum signal amplitude would correspond to the flux enhancers  166  aligning with the positive polarity loops  190 . 
     FIG. 4C shows the net signal output from either of the first and second receiver windings  180  and  182 . This net signal is equal to the sum of the signals output from the positive and negative polarity loops  190  and  192 , i.e., the sum of the signal shown in FIGS. 4A and 4B. The net signal shown in FIG. 4C should ideally be symmetrical around zero, that is, the positive and negative polarity loops  190  and  192  should be exactly balanced to produce a symmetrical output with zero offset. 
     However, a “DC” (position independent) component often appears in the net signal in a real device. This DC component is the offset signal V o . This offset V o  is an extraneous signal component that complicates signal processing and leads to undesirable position measurement errors. This offset has two sources. 
     First, the full amplitude of the transmitter field passes through the first and second receiver windings  180  and  182 . As outlined above, this induces a voltage in each loop  190  and  192 . The induced voltage nominally cancels because the loops  190  and  192  have opposite winding directions. However, to perfectly cancel the induced voltage in the receiver windings requires the positive and negative loops  190  and  192  to be precisely positioned and shaped, for a perfectly balanced result. The tolerances on the balance are critical because the voltages induced directly into the receiver winding loops  180  and  182  by the transmitter winding  176  are much stronger than the modulation in the induced voltage caused by the flux modulators  166 . 
     Second, the spatially modulated field created by the flux modulators also exhibits an average position-independent offset component. That is, the flux modulators  166  within the magnetic field generated by the transmitter winding  176  all create the same polarity spatial modulation in the magnetic field. For example, when flux disrupters are used, the induced eddy current field from the flux modulators has an offset because the flux disrupters within the transmitter field all create a same polarity secondary magnetic field. At the same time, the space between the flux disrupters does not create a secondary magnetic field. 
     Thus, each positive polarity loop  190  and each negative polarity loop  192  of the receiver windings  180  and  182  sees a net magnetic field that varies between a minimum value and a maximum value having the same polarity. The mean value of this function is not balanced around zero, i.e., it has a large nominal offset. Similarly, when flux enhancers are used, the field modulation due to the flux enhancers has an offset because the enhancers within the transmitter winding  176  all create the same field modulation, while the space between the modulators provides no modulation. Each positive and negative polarity loop  190  and  192  of each receiver winding  180  or  182  therefore sees a modulated field that varies between a minimum value and a maximum value having the same polarity. The mean value of this function also has a large nominal offset. 
     A receiver winding having an equal number of similar positive and negative polarity loops  190  and  192  helps eliminate the offset components. However, any imperfection in the balance between the positive and negative polarity loops  190  and  192  allows residual offsets according to the previous description. 
     Both these offset components are expected to be canceled solely by the symmetry between the positive and negative polarity loops  190  and  192  in the first and second windings  180  and  182 . This puts very stringent requirements on the manufacturing precision of the receiver windings  180  and  182 . Experience in manufacturing a transducer indicates it is practically impossible to eliminate this source of error from the induced current position transducer of a conventional caliper. 
     Furthermore, any deviations in the width or pitch of the flux modulators  166  will unbalance the receiver windings  180  or  182  in a way that is independent of the relative position between the PCB  164  and the read head  158 . 
     Any signal component which is independent of the transducer position, such as the aforementioned offset components, is regarded as an extraneous signal to the operation of the transducer. Such extraneous signals complicate the required signal processing circuitry and otherwise lead to errors which compromise the accuracy of the transducer. 
     One proposed solution attempts to reduce the extraneous coupling between the transmitter and receiver windings simply by placing the receiver winding distant from the field produced by the transmitter winding. However, the effectiveness of this technique alone depends on the degree of separation between the transmitter and receiver windings. Hence, this technique contradicts the need for high accuracy linear caliper of compact size. Alternatively, the transmitter field can be confined with magnetically permeable materials so that the effectiveness of a given degree of separation is increased. However, this technique leads to additional complexity, cost, and sensitivity to external fields, in a practical device. 
     Furthermore, the simple winding configurations disclosed in association with these techniques include no means for creating a device with a measuring range significantly exceeding the span of the transmitter and receive windings. In addition, the simple winding configurations provide no means for significantly enhancing the degree of output signal change per unit of displacement for a given measuring range. Thus, the practical measuring resolution of these devices is limited for a given measuring range. 
     The need for a high accuracy inductive linear caliper which rejects both extraneous signal components and external fields, is compact, of simple construction, and capable of high resolution measurement over an extended measuring range without requiring increased fabrication and circuit accuracies, has therefore not been met previously. 
     SUMMARY OF THE INVENTION 
     This invention provides an electronic caliper using an induced current position transducer with improved winding configurations. The improved winding configurations increase the proportion of the useful output signal component relative to extraneous (“offset”) components of the output signal without requiring increased transducer fabrication accuracy. Furthermore, the winding configurations provide means to enhance the degree of output signal change per unit of displacement for a given measuring range. 
     This is accomplished by winding configurations that minimize and nullify the direct coupling between the transmitter and receiver windings while providing enhanced position-dependent coupling between them through a plurality of coupling windings on the scale which interact with a plurality of spatial modulations of the windings. 
     In particular, this invention includes an electronic caliper using a reduced offset induced current position transducer having a scale and a read head that are movable relative to each other along a measuring axis. The read head includes a pair of receiver windings extending along the measuring axis and positioned in a center portion of the read head. The read head further includes a transmitter winding extending along the measuring axis and positioned laterally from the receiver windings in a direction perpendicular to the measuring axis. 
     In a first embodiment of the electronic caliper using the induced current position transducer of this invention, the transmitter winding is divided into a first transmitter loop and a second transmitter loop, with the first transmitter loop placed on one side of the receiver windings and the second transmitter loop placed on the other side of the receiver windings. The magnetic fields created by the first and second loops of the transmitter winding counteract each other in the area of the receiver winding. This minimizes the extraneous effects of any direct coupling from the transmitter winding to the receiver winding. 
     The scale member has a plurality of first coupling loops extending along the measuring axis and interleaved with a plurality of second measuring loops also extending along the measuring axis. The first coupling loops have a first portion aligned with the first transmitter winding and a second portion aligned with the receiver windings. Similarly, the second coupling loops have a first portion aligned with the second transmitter winding and a second portion aligned with the receiver windings. 
     In a second embodiment of the induced current position transducer of this invention, the transmitter has only one loop, which is placed alongside the receiver windings on the read head. The scale member in this case has a plurality of first coupling loops arrayed along the measuring axis and interleaved with a second plurality of coupling loops also arrayed along the measuring axis. Both the first and second coupling loops have a first portion aligned with the transmitter winding and a second portion aligned with the receiver windings. 
     The first and second portions of each first coupling loop are connected in series and are “untwisted”. Thus, the magnetic fields induced in the first and second portions of the first coupling loops have the same polarity. In contrast, the first and second portion of each second coupling loop are connected in series and are “twisted”. In this case, the magnetic fields induced in the first and second portions of the second coupling loops have opposite polarities. This creates an alternating induced magnetic field along the measuring axis in the area under the receiver winding in response to exciting the transmitter winding. 
     These winding configurations substantially eliminate several extraneous signal components, resulting in simplified signal processing and improved transducer accuracy and robustness in an economical design. 
     This invention provides an improved electronic caliper that uses an induced current position transducer with improved winding configurations. This invention uses a transducer with example embodiments that are described in copending U.S. patent application Ser. No. 08/834,432, filed on Apr. 16, 1997, entitled “REDUCED OFFSET HIGH ACCURACY INDUCED CURRENT POSITION TRANSDUCER” which is hereby incorporated by reference in its entirety. 
     These and other features and advantages of this invention are described in or are apparent from the following detailed description of the preferred embodiments. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The preferred embodiments of this invention will be described in detail, with reference to the following figures, wherein: 
     FIG. 1 shows an electronic caliper using an induced current position transducer having undesirable extraneous signal offset components; 
     FIG. 2 is a cross-sectional view of the caliper of FIG. 1; 
     FIG. 3 shows the induced current position transducer of the electronic caliper of FIG. 1; 
     FIG. 4A shows the position-dependent output of the positive polarity loops of FIG. 3; 
     FIG. 4B shows the position-dependent output of the negative polarity loops of FIG. 3; 
     FIG. 4C shows the net position-dependent output of the positive and negative polarity loops of FIG. 3; 
     FIG. 5 shows an electronic caliper of this invention using a reduced offset high accuracy induced current position transducer; 
     FIG. 6 shows a first embodiment of the scale for the reduced offset induced current position transducer of the electronic caliper of this invention; 
     FIG. 7 shows a first embodiment of the read head for the reduced offset induced current position transducer of the electronic caliper of this invention; 
     FIG. 8 shows a second embodiment of the read-head for the reduced offset induced current position transducer of this invention. 
     FIG. 9 shows the signal amplitudes as a function of the relative position of the scale and read-head of FIG. 8; 
     FIG. 10 shows a schematic vector phase diagram for the three phase windings of FIG. 8; 
     FIG. 11A shows a third embodiment of the scale for the reduced offset induced current position transducer of this invention; 
     FIG. 11B shows a first portion of the scale of FIG. 11A in greater detail; 
     FIG. 11C shows a second portion of the scale of FIG. 11A in greater detail; 
     FIG. 11D shows a third embodiment of the read head usable with the scale of FIG. 11A; 
     FIG. 12A shows a cross-sectional view of the first embodiment of the reduced offset induced current position transducer of this invention; 
     FIG. 12B shows a cross-sectional view of the second embodiment of the reduced offset induced current position transducer of this invention; 
     FIG. 13 is a block diagram of the read head shown in FIG.  8  and its associated signal processing circuits; and 
     FIG. 14 is a timing diagram for one of the three channels of the electronic unit shown in FIG.  13 . 
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     As shown in FIG. 5, an inductive caliper  200  includes an elongated beam  202 . The elongated beam  202  is a rigid or semi-rigid bar having a generally rectangular cross section. A groove  206  is formed in an upper surface of the elongated beam  202 . An elongated measuring scale  204  is rigidly bonded to the elongated beam  202  in the groove  206 . The groove  206  is formed in the beam  202  at a depth about equal to the thickness of the scale  204 . Thus, the top surface of the scale  204  is very nearly coplanar with the top edges of beam  202 . 
     A pair of laterally projecting, fixed jaws  208  and  210  are integrally formed near a first end  212  of the beam  202 . A corresponding pair of laterally projecting movable jaws  216  and  218  are formed on a slider assembly  220 . The outside dimensions of an object are measured by placing the object between a pair of engagement surfaces  214  on the jaws  208  and  216 . Similarly, the inside dimensions of an object are measured by placing the jaws  210  and  218  within an object. The engagement surfaces  222  of the jaws  210  and  218  are positioned to contact the surfaces on the object to be measured. 
     The engagement surfaces  222  and  214  are positioned so that when the engagement surfaces  214  of the jaws  208  and  216  are contacting each other, the engagement surfaces  222  of the jaws  210  and  218  are aligned with each other. In this position, the zero position, both the outside and inside dimensions measured by the caliper  200  should be zero. 
     The caliper  200  also includes a depth bar  224  which is attached to the slider assembly  220 . The depth bar  224  projects longitudinally from the beam  202  and terminates at an engagement end  226 . The length of the depth bar  224  is such that the engagement end  226  is flush with a second end  228  of the beam  202  when the caliper  200  is at the zero position. By resting the second end  228  of the beam  202  on a surface in which a hole is formed and extending the depth bar  224  into the hole until the end  226  touches the bottom of the hole, the caliper  200  is able to measure the depth of the hole. 
     Whether a measurement is made using the outside measuring jaws  208  and  216 , the inside measuring jaws  210  and  218 , or the depth bar  224 , the measured dimension is displayed on a conventional digital display  234 , which is mounted in a cover  236  of the caliper  200 . A pair of push button switches  230  and  232  are also mounted in the cover  236 . The switch  230  turns on and off a signal processing and display electronic circuit  260  of the slider assembly  220 . The switch  232  is used to reset the display  234  to zero. 
     As shown in FIG. 5, the slider assembly  220  includes a base  238  with a guiding edge  240 . The guiding edge  240  contacts a side edge  242  of the elongated beam  202  when the slider assembly  220  straddles the elongated beam  202 . This ensures accurate operation of the caliper  200 . A pair of screws  244  bias a resilient pressure bar  246  against a mating edge of the beam  202  to eliminate free play between the slider assembly  220  and the elongated beam  202 . The depth bar  224  is inserted into a depth bar groove  248  formed on an underside of the elongated beam  202 . The depth bar groove  248  extends along the underside of the elongated beam  202  to provide clearance for the depth bar  224 . The depth bar  224  is held in the depth bar groove  248  by an end stop  250 . The end stop  250  is attached to the underside of the beam  202  at the second end  228 . The end stop  250  also prevents the slider assembly  220  from inadvertently disengaging from the elongated beam  202  at the second end  228  during operation. 
     The slider assembly  220  also includes a read head assembly  252  mounted on the base  238  above the elongated beam  202 . Thus, the base  238  and read head assembly  252  move as a unit. The read head assembly  252  includes a substrate  254 , such as a conventional printed circuit board. The substrate  254  bears an inductive read head  258  on its lower surface. A signal processing and display electronic circuit  260  is mounted on an upper surface of the substrate  254 . A resilient seal  256  is compressed between the cover  236  and the substrate  254  to prevent contamination of the signal processing and display electronic circuit  260 . 
     The slider assembly  220  carries the read head  258  so that it is slightly separated from the beam  202  by an air gap  270  formed between the insulative coatings  262  and  268 . The air gap  270  is preferably on the order of 0.5 mm. Together, the read head  258  and the flux couplers  266  form an inductive transducer. 
     FIGS. 6 and 7 show a first embodiment of the reduced-offset incremental induced current position transducer  200  used in the electronic caliper of this invention, which produces an output type usually referred to as “incremental”. “Incremental” output is defined as a cyclic output which is repeated according to a design-related increment of transducer displacement. 
     In particular, FIG. 6 shows a first embodiment of the reduced offset scale  204  of the transducer  200 . The reduced-offset scale  204  includes a first plurality of coupling loops  274  interleaved with a second plurality of coupling loops  276 . Each of the coupling loops  274  and  276  is electrically isolated from the others of the first and second plurality of coupling loops  274  and  276 . 
     Each of the first plurality of coupling loops  274  includes a first loop portion  278  and a second loop portion  280  connected by a pair of connecting conductors  282 . Similarly, each of the second plurality of coupling loops  276  includes a first loop portion  284  and a second loop portion  286  connected by a pair of connecting conductors  288 . 
     In the first plurality of coupling loops  274 , the first loop portions  278  are arranged along one lateral edge of the scale  204  and are arrayed along a measuring axis  272 . The second loop portions  280  are arranged along the center of the scale  204  and are arrayed along the measuring axis  272 . The connecting conductors  282  extend perpendicularly to the measuring axis  272  to connect the first loop portions  278  to the second loop portions  280 . 
     Similarly, in the second plurality of coupling loops  276 , the first loop portions  284  are arranged along a second lateral edge of the scale  204  and arrayed along the measuring axis  272 . The second loop portions  286  are arranged along the center of the scale  204  along the measuring axis  272 , interleaved with the second loop portions  280  of the second coupling loops  276 . The connecting conductors  288  extend generally perpendicularly to the measuring axis  272  to connect the first loop portions  284  to the second loop portions  286 . 
     As shown in FIG. 7, the read head  258  of the transducer  200  includes a transmitter winding  290  having a first transmitter winding portion  292 A and a second transmitter winding portion  292 B. The first transmitter winding portion  292 A is provided at a first lateral edge of the read head  258  while the second transmitter winding portion  292 B is provided at the other lateral edge of the read head  258 . Each of the first and second transmitter winding portions  292 A and  292 B have the same long dimension extending along the measuring axis  272 . Furthermore, each of the first and second transmitter winding portions  292 A and  292 B have a short dimension that extends a distance d 1  in a direction perpendicular to the measuring axis  272 . 
     The terminals  290 A and  290 B of the transmitter winding  290  are connected to the transmitter drive signal generator  294 . The transmitter drive signal generator  294  outputs a time-varying drive signal to the transmitter winding terminal  292 A. Thus a time-varying current flows through the transmitter winding  292  from the transmitter winding terminal  292 A to the transmitter winding terminal  292 B. 
     In response, the first transmitter winding portion  292 A generates a magnetic field that rises up out of the plane of FIG. 7 inside the first transmitter winding portion  292 A and descends into the plane of FIG. 7 outside the loop formed by the first transmitter winding portion  292 A. In contrast, the second transmitter winding portion  292 B generates a primary magnetic field that rises up out of the plane of FIG. 7 outside the loop formed by the second transmitter winding portion  292 B and descends into the plane of FIG. 7 inside the loop formed by the second transmitter winding portion  292 B. 
     A current is then induced in the coupling loops  274  and  276  that counteracts the change of magnetic field. Thus, the induced current in each of the coupling loop sections  278  and  284  flows in a direction opposite to the current flowing in the respective adjacent portions of the transmitter loops  292 A and  292 B. As shown in FIG. 7 adjacent ones of the second loop portions  280  and  286  in the center section of the scale have loop currents having opposite polarities. Thus, a secondary magnetic field is created having field portions of opposite polarity periodically distributed along the center section of the scale. The wavelength λ of the periodic secondary magnetic field is equal to the distance between successive second loop portions  280  (or  286 ). 
     The read head  258  also includes first and second receiver windings  296  and  298  that are generally identical to the first and second receiver windings  180  and  182  shown in FIG.  3 . In particular, similarly to the first and second receiver windings  180  and  182  shown in FIG. 3, the first and second receiver windings  296  and  298  are each formed by a plurality of sinusoidally-shaped loop segments  300  and  302  formed on opposite sides of an insulating layer of the printed circuit board forming the read head  258 . 
     The loop segments  300  and  302  are linked through feed-throughs  304  to form alternating positive polarity loops  306  and negative polarity loops  308  in each of the first and second receiver windings  296  and  298 . The receiver windings  296  and  298  are positioned in the center of the read head  258  between the first and second transmitter portions  292 A and  292 B. Each of the first and second receiver windings  296  and  298  extends a distance d 2  in the direction perpendicular to the measuring axis. 
     Extraneous (position independent and scale independent) coupling from the transmitter loops to the receiver loops is generally avoided in this configuration. That is, the primary magnetic fields generated by the first and second transmitter portions  292 A and  292 B point in opposite directions in the vicinity of the first and second receiver windings  296  and  298 . Thus, the primary magnetic fields counteract each other in the area occupied by the first and second receiver windings  296  and  298 . Ideally, the primary magnetic fields completely counteract each in this area. The first and second receiver windings  296  and  298  are spaced equal distances d 3  from the inner portions of the first and second transmitter winding portions  292 A and  292 B. Thus, the magnetic fields generated by each of the first and second transmitter winding portions  292 A and  292 B in the portion of the read head  258  occupied by the first and second receiver windings  296  and  298  are in symmetric opposition and the associated inductive effects effectively cancel each other out. The net voltage induced in the first and second receiver windings  296  and  298  by extraneous direct coupling to the first and second transmitter winding portions  292 A and  292 B is reduced to a first extent by positioning the transmitter windings away from the receiver windings. Secondly, the symmetric design effectively reduces the net extraneous coupling to zero. 
     Each of the first plurality of coupling loops  274  is arranged at a pitch equal to a wavelength λ of the first and second receiver windings  296  and  298 . Furthermore, the first loop portions  278  each extends a distance along the measuring axis  272  which is as close as possible to the wavelength λ while still providing an insulating space  310  between adjacent ones of the first loop portions  276  and  278 . In addition, the first loop portions  276  and  278  extend the distance d 1  in the direction perpendicular to the measuring axis  272 . 
     Similarly, the second plurality of coupling loops  276  are also arranged at a pitch equal to the wavelength λ. The first loop portions  284  also extend as close as possible to each other along the measuring axis to the wavelength λ while providing the space  310  between adjacent ones of the first loop portions  284 . The first loop portions  284  also extend the distance d 1  in the direction perpendicular to the measuring axis  272 . 
     The second loop portions  280  and  286  of the first and second pluralities of coupling loops  274  and  276  are also arranged at a pitch equal to the wavelength λ. However, each of the second loop portions  280  and  286  extends along the measuring axis as close as possible to only one-half the wavelength λ. An insulating space  312  is provided between each adjacent pair of second loop portions  280  and  286  of the first and second pluralities of coupling loops  274  and  276 , as shown in FIG.  7 . Thus, the second loop portions  280  and  286  of the first and second pluralities of coupling loops  274  and  276  are interleaved along the length of the scale  204 . Finally, each of the second loop portions  280  and  286  extends the distance d 2  in the direction perpendicular to the measuring axis  272 . 
     As shown in FIG. 7, the second loop portions  280  and  286  are spaced the distance d 3  from the corresponding first loop portions  278  and  284 . Accordingly, when the read head  258  is placed in proximity to the scale  204 , as shown in FIG. 7, the first transmitter winding portion  292 A aligns with the first loop portions  278  of the first plurality of coupling loops  274 . Similarly, the second transmitter winding portion  292 B aligns with the first loop portions  284  of the second plurality of coupling loops  276 . Finally, the first and second receiver windings  296  and  298  align with the second loop portions  280  and  286  of the first and second coupling loops  274  and  276 . As will be apparent from the preceding and the following discussions, the area enclosed by the second loop portions  280  and  286  define a sensing track extending parallel to the measuring axis, and that substantially all of the effective magnetic field passing through the sensing track is due solely to the current flow in the second loop portions. 
     In operation, a time-varying drive signal is output by the transmitter drive signal generator  294  to the transmitter winding terminal  290 A. Thus, the first transmitter winding portion  292 A generates a first changing magnetic field having a first direction while the second transmitter winding portion  292 B generates a second magnetic field in a second direction that is opposite to the first direction. This second magnetic field has a field strength that is equal to a field strength of the first magnetic field generated by the first transmitter winding portion  292 A. 
     Each of the first plurality of coupling loops  274  is inductively coupled to the first transmitter winding portion  292 A by the first magnetic field generated by the first transmitter winding portion  292 A. Thus, an induced current flows clockwise through each of the first plurality of coupling loops  274 . At the same time the second plurality of coupling loops  276  is inductively coupled to the second transmitter winding portion  292 B by the second magnetic field generated by the second transmitter winding portion  292 B. This induces a counterclockwise current to flow in each of the second plurality of coupling loops  276 . That is, the currents through the second portions  280  and  286  of the coupling loops  274  and  276  flow in opposite directions. 
     The clockwise flowing current in each of the second portions  280  of the first coupling loops  274  generates a third magnetic field that depends into the plane of FIG. 7 within the second portions  280 . In contrast, the counterclockwise flowing currents in the second loop portions  286  of the second coupling loops  276  generate a fourth magnetic field that rises out of the plane of FIG. 7 within the second loop portions  286  of the second coupling loops  276 . Thus, a net alternating magnetic field is formed along the measuring axis  272 . This net alternating magnetic field has a wavelength which is equal to the wavelength λ of the first and second receiver windings  296  and  298 . 
     Accordingly, when the positive polarity loops  306  of the first receiver winding  296  are aligned with either the second loop portions  280  or  286 , the negative polarity loops  308  of the first receiver winding  296  are aligned with the other of the second loop portions  280  or  286 . This is also true when the positive polarity loops  306  and the negative polarity loops  308  of the second receiver winding  298  are aligned with the second loop portions  280  and  286 . Because the alternating magnetic field generated by the second loop portions  280  and  286  is spatially modulated at the same wavelength as the spatial modulation of the first and second receiver windings  296  and  298 , the EMF generated in each of the positive and negative polarity loops  306  and  308  when aligned with the second loop portions  280  is equal and opposite to the EMF generated when they are aligned with the second loop portions  286 . 
     Thus, the net output of the positive polarity loops  306 , as the read head  258  moves relative to the scale  204  is a sinusoidal function of the relative position of the read head along the scale and the offset component of the output signal due to extraneous coupling is nominally zero. Similarly, the net output from the negative polarity loops  308 , as the read head  258  moves relative to the scale  204 , is also sinusoidal and centered on the position axis. The EMF output from the positive polarity loops  306  and the negative polarity loops  308  are in phase. They thus generate a net position-dependent output signal corresponding to FIG. 4C, but without the DC offset V o . 
     Finally, the first and second receiver windings  296  and  298 , like the first and second receiver windings  138  and  140 , are in quadrature. Thus, the output signal generated by the first receiver winding  296  and output to the receiver signal processing circuit  314  is 90 degrees out of phase with the signal output by the second receiver winding  298  to the receiver signal processing circuit  314 . 
     The receiver signal processing circuit  314  inputs and samples the output signals from the first and second receiver windings  296  and  298 , converts the signals to digital values and outputs them to the control unit  316 . The control unit  316  processes these digitized output signals to determine the relative position between the read head  258  and the scale  204  within a wavelength λ. 
     It should be appreciated that, with a suitable feed-through arrangement, either the positive polarity loops  306  or the negative polarity loops  308  could be reduced to zero width perpendicular to the measuring axis (becoming effectively simple conducting elements between the adjacent loops). In this case, the first and second receiver windings  296  and  298  become unipolar flux receivers, introducing an increased sensitivity to external fields, and reducing their output signal amplitude to half that of the previously described embodiment (due to the eliminated loop area). 
     However, the modified design retains many inventive benefits. The net extraneous flux through the loops is still nominally zero due to the symmetric transmitter winding configuration. The output signal from each receiver winding  296  and  298  still swings from a maximum positive value to a maximum negative value with nominally zero offset. The degree of output signal change per unit of displacement, for a given measuring range, is still very high, due to the complementary periodic structure of the scale elements and receiver windings. 
     Based on the nature of the quadrature output from the first and second receiver windings  296  and  298 , the control unit  316  is able to determine the direction of relative motion between the read head  258  and the scale  204 . The control unit  316  counts the number of partial or full “incremental” wavelengths λ traversed, by signal processing methods well-known to those skilled in the art and disclosed herein and in the incorporated references. The control unit  316  uses that number and the relative position within a wavelength λ to output the relative position between the read head  258  and the scale  204  from a set origin. 
     The control unit  316  also outputs control signals to the transmitter drive signal generator  294  to generate the time-varying transmitter drive signal. It should be appreciated that any of the signal generating and processing circuits shown in U.S. patent application Ser. No. 08/441,769, filed May 16, 1995, U.S. patent application Ser. No. 08/645,483, filed May 13, 1996 and U.S. patent application Ser. No. 08/788,469, filed Jan. 29, 1997 hereby incorporated by reference, can be used to implement the receiver signal processing circuit  314 , the transmitter drive signal generator  294  and the control unit  316 . Thus, these circuits will not be described in further detail herein. 
     FIG. 8 shows a second embodiment of a read head that can be used with a scale according to FIG.  6 . The receiver in this version of the read head has three receiver windings  318 ,  320  and  322 . The receiver windings are offset from each other along the measurement axis by ⅓ of the wavelength λ. FIG. 9 shows the signal functions from the three receivers as a function of the position along the measurement axis. 
     It should be appreciated that perfectly sinusoidal output functions are difficult to achieve in practice, and that deviations from a perfect sinusoidal output contain spatial harmonics of the fundamental wavelength of the transducer. Therefore, the three phase configuration of this second embodiment of the reduced-offset induced current position transducer has a significant advantage over the first embodiment of the reduced offset induced current position transducer, in that the third harmonic content in the separate receiver windings&#39; signal can be largely eliminated as a source of position measurement error. 
     Eliminating the third harmonic is accomplished by combining the outputs of the receiver windings as shown in FIG. 10, where the three windings are connected in a star configuration and the signals used for determining position are taken between the corners of the star. This can also be accomplished by measuring each of the output signals independently from the receiver windings  318 ,  320  and  322 , and then combining them computationally in a corresponding way in a digital signal processing circuit. The following equations outline how the third harmonic component is eliminated by suitably combining the original three phase signals, designated as U R , U S , and U T . 
     Assume each of the unprocessed phase signals contains the fundamental sinusoidal signal plus the third harmonic signal, with equal amplitude in the three phases, then:                U   R     =                    A   0        sin                   (     2      π                   x   λ       )       +       A   3        sin                   (     2      π                     3      x     λ       )                       U   S     =                      A   0     ·   sin                     (     2      π          x   +     λ   3       λ       )       +       A   3        sin                   (     2      π          3                   (     x   +     λ   3       )       λ       )                     =                      A   0     ·   sin                     (       2      π        x   λ       +       2      π     3       )       +       A   3        sin                   (       2      π          3      x     λ       +     2      π       )                       =                      A   0     ·   sin                     (       2      π                   x   λ       +       2                 π     3       )       +       A   3        sin                   (     2      π          3      x     λ       )           ;                 U   T     =                      A   0     ·   sin                     (     2      π          x   -     λ   3       λ       )       +       A   3        sin                   (     2      π          3                   (     x   -     λ   3       )       λ       )                     =                      A   0     ·   sin                     (       2      π        x   λ       -       2      π     3       )       +       A   3        sin                   (       2      π          3      x     λ       -     2      π       )                     =                      A   0     ·   sin                     (       2      π                   x   λ       -       2                 π     3       )       +       A   3        sin                   (     2      π          3      x     λ       )                               
     Creating new signals by pair-wise subtracting the above-outlined signals from each other eliminates the third harmonic to provide:                      V   R     =         U   T     -     U   S       =         A     0                       (       sin                   (       2      π        x   λ       -       2      π     s       )       -     sin                   (       2      π        x   λ       +       2      π     3       )         )       =       -     A   0            3        cos                 2      π        x   λ                         V   S     =         U   R     -     U   T       =         A     0                       (       sin                   (     2      π        x   λ       )       -     sin                   (       2      π        x   λ       -       2      π     3       )         )       =       A   0          3        cos                   (       2      π        x   λ       -       2      π     6       )                               V   T     =         U   S     -     U   R       =         A     0                       (       sin                   (       2      π                   x   λ       +       2      π     3       )       -     sin                   (     2      π                   x   λ       )         )       =       A   0          3        cos                   (       2      π        x   λ       +       2      π     6       )                                 
     To get quadrature signals for position calculation in the same way, V S  and V T  are combined:                V   Q     =         V   S     -     V   T       =                  A   0          3                     (       cos                   (       2      π                   x   λ       -       2      π     6       )       -     cos                   (       2      π                   x   λ       +       2      π     6       )         )                     =                    A   0          3     *   2      sin                 2      π                   x   λ        sin                   (     -       2      π     6       )       =       A   0        3                 sin                 2      π                   x   λ                               
     After identifying the applicable quarter-wavelength position quadrant within the incremental wavelength, the interpolated position within the quarter wavelength is then calculated by:            V   Q       -     V   R         =       3     *   tan                   (     2      π                   x   λ       )                       
     Solving for x:        x   =       λ     2      π       *     tan   1                     (       V   Q         -     V   R       *     3         )                       
     The position calculated this way using the output from three phase receiver windings will not contain any error from third harmonic components in the receiver output signal functions, to the extent that the outputs from all three receiver windings have the same third harmonic characteristics, which is generally the case for practical devices. Also, if the receiver signals are amplified in preamplifiers in the electronic unit, the measurement error caused by certain distortion errors in those electronic preamplifiers will be canceled by the above described signal processing in the three phase configuration. 
     FIGS. 11A-11D show a third embodiment of the read head and scale for the reduced offset induced current position transducer of the linear scale of this invention. This embodiment contains only one transmitter winding loop  490 , which is placed on one side of the receiver windings  496  and  498  on the read head  458 . The scale  404  is a two layer printed circuit board (PCB). Pattern forming coupling loops  474  and  476  are arrayed on the scale  404  along the measurement axis. 
     Each coupling loop  474  includes a first loop portion  478  which is connected by connection lines  482  to a second loop portion  480 . The first and second loop portions  478  and  480  are connected so that an induced current produces the same polarity field in the first loop portion  478  and the second loop portion  480 . Each coupling loop  476  includes a first loop portion  484  which is connected by connection lines  488  to a second loop portion  486 . The first and second loop portions  484  and  486  are connected so that an induced current produces fields having opposite polarities in the first and second loop portions  484  and  486 . 
     The detailed construction of the coupling loops  474  and  476  is shown in FIGS. 11B and 11C. FIG. 11B shows a first conductor pattern provided on a first one of the layers of the PCB forming the scale  404 . FIG. 11C shows a second construction pattern provided on a second one of the layers of the PCB forming the scale  404 . The individual portions of the first and second patterns formed on the first and second layers are connected via the feed-throughs  504  of the PCB to form the coupling loops  474  and  476 . 
     The read head  458  is formed by a second PCB and includes a transmitter loop  490  and first and second receiver windings  496  and  498 . The first and second receiver windings  496  and  498  are in this embodiment in a two-phase configuration. This embodiment could also use the three-phase configuration previously disclosed. The transmitter loop  490  encloses an area that covers the first loop portions  478  and  484  over the length of the read head. The transmitter loop  490  is excited in the same way as described previously in conjunction with FIG.  7 . 
     The first loop portions  478  and  484  of the coupling loops  474  and  476  under the transmitter loop  490  respond to the primary magnetic field generated by the transmitter  490  with an induced EMF that causes a current and magnetic field that counteracts the primary magnetic field produced in the transmitter winding  490 . When the transmitter winding current flows counter-clockwise, as shown in FIG. 11D, the induced current in the first loop portions  478  and  484  of the coupling loops  474  and  476  flows counterclockwise. The current in the second loop portions  480  of the coupling loops  474  also flows clockwise. However, the current in the second loop portions  486  of the coupling loops  476  flow counter-clockwise because of the crossed connections  488  described above. 
     Therefore, the array of second loop portions  480  and  486  produces a secondary magnetic field with regions of opposite polarity periodically repeating along the scale under the receiver windings  496  and  498  of the read head unit  458 . The secondary magnetic field has a wavelength λ equal to the period length for successive ones of the second loop portions  480 , which is also equal to the period length for successive ones of the second loop portions  486 . The receiver loops of the first and second windings  496  and  498  are designed to have the same wavelength λ as the scale pattern. 
     Hence, the receiver loops of the first and second receiver windings  496  and  498  will exhibit an induced EMF which produces a signal voltage whose amplitude will follow a periodic function with wavelength λ when the read head  458  is moved along the scale  404 . Thus, except for the distinction of the single transmitter loop  490 , this embodiment functions in the manner previously described for the embodiment shown in FIGS. 6 and 7. Similar to the previous discussion of second loop portions  280  and  286  of FIG. 7, the total area enclosed by the second loop portions  480  and  486  define a sensing track extending parallel to the measuring axis. In this case, the effective magnetic field within the sensing track includes some effect due to coupling to the fringe of the field produced by the transmitter winding  490 . However, the current flow in the second loop portions produces a field in the sensing track that predominates over any other field. 
     FIG. 12A shows a cross-section of an inductive read head according to the first embodiment of this invention shown in FIG.  7 . FIG. 12A illustrates how the primary magnetic field caused by the current in the transmitter loop  292 A encircles the conductors and partly crosses through the receiver loops  296  and  298 . FIG. 12A also shows how the primary magnetic field caused by the current in the transmitter loop  292 B passes through the receiver loops  296  and  298  in the opposite direction from the primary magnetic field caused by the transmitter loop  292 A. 
     Thus, the resulting net magnetic field through the first and second receiver windings  296  and  298  will be very close to zero and the extraneous direct coupling from the transmitter loops  292 A and  292 B to the first and second receiver windings  296  and  298  will be nullified. Experience and theoretical calculations show an improvement in the ratio of useful to extraneous signal components by a factor of more than 100 relative to the embodiment shown in FIG.  3 . 
     FIG. 12B shows a cross-section of an inductive read head according to the third embodiment of this invention shown in FIG.  11 D. FIG. 12B illustrates how the primary magnetic field caused by the current in the transmitter loop  490  encircles the conductors and partly crosses through the first and second receiver loops  496  and  498 . Although this case fails to nullify the extraneous direct coupling, as provided in the first preferred embodiment, it still reduces the extraneous direct coupling by virtue of the separation of the transmitter loop  490  and the first and second receiver windings  496  and  498 . 
     Furthermore, the secondary magnetic field having alternating polarities is provided in the vicinity of the first and second receiver windings  496  and  498 . This eliminates other sources of offset. According to experience and theoretical calculations, the third embodiment shows an improvement in the ratio of useful to extraneous signal components by a factor of about 10 relative to the embodiment shown in FIG.  3 . 
     It should be appreciated that the previous embodiments may be modified in certain aspects, while retaining many of their inventive benefits. For example, the coupling loops  474  (or  476 ) of FIG. 11A may be eliminated, while other aspects of this configuration remain the same. In this case, the secondary magnetic field provided in the vicinity of the first and second receiver windings  496  and  498  does not have a pattern of alternating polarities, as in the third embodiment. However, this design still reduces the extraneous direct coupling between transmitter and receiver windings by virtue of the separation of the transmitter loop  490  and the first and second receiver windings  496  and  498 . 
     Furthermore, the use of multiple coupling loops provides the benefit of averaging out the error contributions of small, but significant, random deviations in segments of the winding configurations due to imperfect fabrication processes. Also, even if the coupling loops  474  (or  476 ) are eliminated, the fundamental operation of the transducer is still based on a moving structured field, defined by the coupling loops  474  (or  476 ) providing the primary excitation for the first and second receiver windings  496  and  498 . It should also be noted that the vertical sections of the first loop portions  478  and  484  shown in FIG. 11D could be bridged by horizontal conductors at the top and bottom (not shown). In this case, the multiple coupling loops form a single coupling loop with a single elongated portion under the transmitter winding  490 , and multiple serially connected loop portions  480  and  486  under the windings  496  and  498 . Thus, the moving structured field is still maintained, although the function of the first coupling loop portions  478  and  484  is now provided by a single continuous winding. 
     In contrast, in the embodiment shown in FIG. 3, a spatially static uniform field provided the primary excitation for the first and second receiver windings  180  and  182 . The receiver winding output signals are based on how this uniform field is affected by moving elements which disturb the uniform excitation field in the vicinity of the first and second receiver windings  180  and  182 . The moving structured field excitation approach of this invention provides an inherently superior signal, even if the coupling loops  474  (or  476 ) are eliminated. 
     FIG. 13 shows a block diagram of the second embodiment of the reduced offset induced current position transducer  200  using the three phase read head  258  shown in FIG.  8 . Only the essential portions of the signal processing circuit needed to determine the position of the read head  258  relative to the scale  204  are shown in FIG.  13 . 
     As shown in FIG. 13, the transmitter winding  290  is connected to a signal generator circuit  295  of the transmitter drive signal generator  294 . The signal generator circuit  295  includes a first switch  324  serially connected to a second switch  326  between ground and a power supply voltage V DD  from an energy source  328 . One terminal of a capacitor  330  is connected to a node N 1  between the first and second switches  324  and  326 . A second plate of the capacitor  330  is connected to the terminal  290 A of the transmitter winding  290 . The second terminal  290 B of the transmitter winding  290  is connected to ground. This, the transmitter winding  290  forms the inductor in a LC resonant circuit with the capacitor  330 . 
     The transmitter winding  290  is indirectly inductively coupled via the coupling loops  274  and  276  formed on the scale  204  to the first-third receiver windings  318 ,  320  and  322 . The receiver windings  318 ,  320  and  322  are connected to a sample and hold circuit  332 . In particular, the output of the first receiver  318  is connected to a first sample and hold subcircuit  334 . The output of the second receiver  320  is connected to a second sample and hold subcircuit  336 , while the output of the third receiver  322  is connected to a third sample and hold subcircuit  338 . 
     Each of the three sample and hold subcircuits  334 ,  336  and  338  includes a switch  340  receiving an output from the corresponding receiver loop  318 ,  320 , or  322 . The output of the switch  340  is connected to the positive input terminal of a buffer amplifier  342 . One plate of a sample and hold capacitor  344  is connected to a node N 3  between the switch  340  and the buffer amplifier  342 . The other plate of the sample and hold capacitor  344  is connected to ground. An output of the buffer amplifier  342  is connected to a switch  346 . The negative input terminal of the buffer amplifier  342  is connected to the output of the buffer amplifier at a node N 4 . 
     The outputs of the switches  346  of the three sample and hold subcircuits  334 ,  336  and  338  are connected to a single output line  348  that is connected to an input of analog-to-digital (A/D) converter  350 . The A/D converter  350  converts the output of the sample and hold circuit  332  from an analog value to a digital value. The digital value is output to a microprocessor  352  which processes the digital values from the A/D converter to determine the relative position between the read head  258  and the scale  204 . 
     Each position within a wavelength can be uniquely identified by the microprocessor  352 , according to known techniques and the equations previously disclosed herein. The microprocessor  352  also uses known techniques to keep track of the direction of motion and the number of wavelengths that are traversed to determine the position for the transducer relative to an initial reference position. 
     The microprocessor  352  also controls the sequence of signal sampling by outputting a control signal over a signal line  354  to a digital control unit  356 . The digital control unit  356  controls the sequence of transmission, signal sampling and A/D conversion by outputting control signals on the signal lines  358 ,  360 ,  362 ,  364 ,  366  and  368  to the transmitter drive signal generator  294  and the sample and hold circuit  332 . In particular, as shown in FIG. 13, the digital control unit  356  outputs switch control signals over the signal lines  358  and  360  to the first and second switches  324  and  326 , respectively, for controlling the transmitter excitation. 
     The digital control unit  356  outputs switch control signals on the signal lines  362 ,  364 ,  366  and  368  to the sample and hold circuit  332 . In particular, the control signal  362  controllably opens and closes the switches  340  of the first-third sample and hold subcircuits  334 ,  336  and  338  to connect the receiver windings  318 ,  320  and  322  to the sample and hold capacitors  344 . When the control signal  362  controllably opens the switches  340 , the signals received from the receiver windings  318 ,  320  and  322  are stored in the sample and hold capacitors  344 . The switch control signals on the signal lines  364 ,  366  and  368  are used to controllably connect the outputs of the buffer amplifiers  342  of one of the first-third sample and hold subcircuits  334 ,  336 , and  338 , respectively, to the A/D converter  350  over the signal line  348 . 
     FIG. 14 shows a timing diagram for generating the switch control signals  358 ,  360 ,  362 ,  364 ,  366  and  368  to obtain a position measurement. First, the switch control signal output on the signal lines  358  is set to a high state to close the switch  324 . This charges up the capacitor  330  to the supply voltage V DD . The switch control signal on the signal line  358  is then set to a low state to open the switch  324 . 
     Next, the switch control signal output on the signal line  360  is changed from a low state to a high state to close the switch  326 . This allows the capacitor  330  to discharge through the corresponding transmitter winding  290 . In particular, the capacitor  330  forms a resonant circuit with the transmitter windings  290  with a chosen resonant frequency on the order of several MHz. The resonance is a damped oscillation with a waveform corresponding essentially to the signal S X  shown in FIG.  14 . 
     The signal S X  appears with the same time function on each of the receiver windings  318 ,  320  and  322 . However, the amplitude and polarity of the signal S X  appearing on each of the receiver windings  318 ,  320  and  322  depends on the position of the read head  250  relative to the scale  204 , as shown in FIG.  9 . 
     Before the signal S X  on the receiver windings reaches a peak, the switch control signal on the signal line  362  changes from a low state to a high state to begin charging each of the sample and hold capacitors  344  of the sample and hold circuit  332 . At a point just after, but approximately at, the peak of the signal S X , the switch control signal on the signal line  362  returns to the low state to open the switches  340 . This holds the amplitude of the signals S X  for each of the three receiver windings on the corresponding one of the sample and hold capacitors  344  of the first-third sample and hold subcircuits  334 ,  336  and  338 . At some point thereafter, the switch control signal on the signal line  360  is returned to the low state to open the switch  326 . 
     Next, at some time after the control signal  362  has returned to the low state, the switch control signal on the signal line  364  changes from the low state to the high state to close the switch  346  of the sample and hold subcircuit  334 . This connects the sampled value held on the corresponding sample and hold capacitor  344  over the signal line  348  to the A/D converter  350 . The A/D converter  350  converts the analog value on the signal line  348  to a digital value and outputs the digital value to the microprocessor  352 . The switch control signal on the signal line  364  returns to the low state to open the corresponding switch  346 . This sequence is then repeated for the switch control signals output on the signal lines  366  and  368  to connect the signals sampled by the sample and hold subcircuits  336  and  338  to the A/D converter  350  over the signal line  348 . 
     This process is repeated according to the program in the microprocessor. A program can easily be made that adapts the sampling rate of the system to the speed of movement of the transducer, thereby minimizing the current consumption. This operation is well known to those skilled in the art and thus will not be described in further detail herein. 
     The previously described signal processing system can be operated on very low power with the disclosed inductive position transducers, and other related inductive position transducers, if desired. For example, intermittently activating the drive signal generator  295  to support a signal processing system sampling frequency of about 1 kHz provides sufficient accuracy and motion tracking capability for most applications. To reduce power consumption, the drive signal generator duty cycle can be kept low by making the pulses relatively short. For example, for the 1 kHz sampling frequency described above, a suitable pulse width is about 0.1-1.0 μs. That is, the duty cycle of the pulses having sampling period of 1 ms is 0.01%-0.1%. 
     The resonant frequency of the capacitor  330  and the winding  290  is then preferably selected such that the peak of the voltage across the capacitor  330  occurs before the end of the 1.0 μs or less pulse. Thus, the resonant frequency is on the order of several megahertz, as previously disclosed. The corresponding magnetic flux will therefore be modulated at a frequency above 1 MHz, and typically of several megahertz. This is considerably higher than the frequencies of conventional inductive position transducers. 
     The inventors have determined that, at these frequencies, the currents generated in the scale  204  with the coupling loops  274  and  276  produce strong inductive coupling to the first-third receiver windings  318 ,  320  and  322 . The EMFs generated in the first-third receiver windings  318 ,  320  and  322 , and the resulting output signal, therefore respond strongly to variations in coupling loop position. This occurs despite the low duty cycle and low power used by the pulsed drive signal. 
     The strength of the response, combined with the low duty cycle and low power consumption, allows the inductive position transducer to make measurements while the drive signal generator  294  and the remainder of the signal processing electronic circuit shown in FIG. 13 draw an average current preferably below 200 μA, and more preferably below 75 μA, for lower power applications. It should be understood that “average current” as used herein means the total charge consumed over one or more measurement cycles, divided by the duration of the one or more measurement cycles, while the inductive position transducer is in normal use. 
     The inductive position transducers similar to the type disclosed herein can therefore be operated with an adequate battery lifetime using three or fewer commercially available miniature batteries or with a photo-electric cell. Further details regarding low power signal processing are disclosed in the incorporated references. 
     It should be appreciated that although the foregoing embodiments are shown with spatially uniform windings designated as the transmitter windings, and spatially modulated windings designated as the receiver windings, it will be apparent to one skilled in the art that the disclosed transducer winding configurations will retain all of their inventive benefits if the roles of the transmitter and receiver windings are “reversed” in conjunction with appropriate signal processing. One such appropriate signal processing technique is disclosed in reference to FIG. 21 of incorporated U.S. patent application Ser. No. 08/441,769. Other applicable signal processing techniques will be apparent to those skilled in the art. 
     Thus, while this invention has been described in conjunction with the specific embodiments outlined above, it is evident that many alternatives, modifications and variations will be apparent to those skilled in the art. Accordingly, the preferred embodiments of the invention as set forth above are intended to be illustrative, not limiting. Various changes may be made without departing from the spirit and scope of the invention as defined in the following claims.