Patent Publication Number: US-2020304049-A1

Title: Pulse width modulation pattern generator and corresponding systems, methods and computer programs

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims priority to International Application No. PCT/CN2018/112206 filed on Oct. 26, 2018, which is incorporated by reference herein in its entirety. 
    
    
     TECHNICAL FIELD 
     The present application relates to pulse width modulation (PWM) pattern generators and corresponding systems, methods and computer programs. 
     BACKGROUND 
     Permanent magnet synchronous motors (PMSMs) are used in a variety of applications, including automotive, industrial and consumer applications. For hybrid electrical vehicles and electrical vehicles, like electrical cars, PMSMs are used, e.g., as motor generators both to drive the vehicle and to generate current for the vehicle for example during deceleration phases. When the motor generator is used as a motor, field-oriented control (FOC) via space vector pulse width modulation (SVPWM) is an often-used approach for driving the motor via a three-phase power inverter. Field oriented control is for example described in U.S. Pat. No. 9,614,473 B1. Also in other applications, an electric motor may be driven using FOC. A three-phase power inverter in many applications includes three half-bridges, each half-bridge comprising two switches like insulated gate bipolar transistors (IGBTs) or other transistors. Such switches are also referred to as power switches. Each half-bridge further comprises two diodes and each diode is coupled in anti-parallel to an associated switch. In anti-parallel means that a forward direction of the diode is opposite to a preferred current flow direction of the associated switch, for example opposite a forward direction of an IGBT used as a switch. These diodes in some switch implementations may be inherent in the design of the switch, whereas in other applications they may be provided separately. Such diodes are also referred to as freewheeling diodes in some contexts. The switches and diodes will be jointly referred to as power devices herein. 
     In operation, when the motor is turning the switches are controlled based on a feedback signal from the motor indicating the angular position using control vectors, or, in other words, a feedback angle. In such a control scheme, the power devices take turns in conducting current flowing through windings of the motor to provide torque for driving the motor. 
     However, this approach may cause problems when the rotor of the motor is locked, i.e., not moving. This may for example occur in certain drive situations in an electric vehicle. In this case, the current always flows through the same power devices determined by the position in which the rotor is locked, which may cause overheating of these power devices, also referred to as hotspots. Similar problems may occur in other cases, e.g., at very slow rotation speeds of the rotor. 
     To further illustrate this, there are three worst case scenarios for electrical vehicles for the operation of a three-phase inverter, which are referred to as the peak power case, the peak torque case and the locked rotor torque case. Peak power often occurs at an acceleration stage, i.e., when the vehicle is accelerated and requires maximum power for acceleration, such that the motor may draw maximum power. The peak torque case occurs for example when driving upward a hill. The locked rotor torque case may occur when starting to drive upwards a hill or climbing an obstacle, i.e., when the angular rotation of the motor of the electrical vehicle is substantially reduces or completely stopped. 
     Generally, the output torque of a motor is proportional to the phase current flowing through the motor. In many designs, the torque in the locked rotor torque case, i.e., the torque generated by the motor in case of a locked rotor, is designed to be close to the peak torque. Since in such designs the power loss at the locked rotor torque is higher than the power loss at peak torque and peak power cases, the locked rotor torque case in such designs may be seen as the worst case. This means that the power loss at the locked rotor torque case determines the design of the power switches when designing the three-phase power inverter, as the power switches have to be able to withstand the hotspot temperature and the power losses in the locked rotor case (e.g., heating due to the power losses). Designing power switches for higher power losses, while possible, generally increases area requirement and the cost of the power switches. 
     SUMMARY 
     According to an embodiment, a system includes pulse width modulation pattern generator configured to be coupled to a three-phase power inverter, wherein the three-phase power inverter comprises three half-bridges, and each half-bridge of the three half-bridges comprises two switches and two diodes coupled in anti-parallel to the switches as power devices, wherein: the pulse width modulation pattern generator is configured to control the three-phase power inverter using field-oriented control via space vector pulse width modulation, in at least one mode of operation, in each control period of the space vector pulse width modulation, at least four of the power devices of the three-phase power inverter take turns in bearing a full current during application of a null vector, the null vector is a vector in which all three half-bridges are controlled to be in a same state, and the full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     According to another embodiment, a method for controlling a three-phase power inverter comprising three half-bridges that each comprise two switches and two diodes coupled in anti-parallel to the switches as power devices, the method comprising: controlling the three-phase power inverter using field-oriented control via space vector pulse width modulation; wherein in at least one mode of operation, in each control period of the space vector pulse width modulation, four of the power devices take turns in bearing a full current during application of a null vector, the null vector is a vector in which all three half-bridges are controlled to be in a same state, and the full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     The above summary is merely intended to give a brief overview over some features of some embodiments and is not to be construed as limiting, as other embodiments may comprise other features than the ones explicitly defined above. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a diagram illustrating a system to an embodiment; 
         FIG. 2  is a flowchart illustrating a method according to an embodiment; 
         FIG. 3  is a diagram illustrating field-oriented control using space vector pulse width modulation; 
         FIG. 4  is a further diagram illustrating field-oriented control using space vector pulse width modulation; 
         FIG. 5  is a diagram of a reference example illustrating conventional field-oriented control; 
         FIG. 6  is a diagram of a further reference example illustrating conventional field-oriented control at another rotor position; 
         FIG. 7  is a diagram illustrating which power devices carry full current in which sector of conventional field-oriented control; 
         FIG. 8  is a diagram illustrating field-oriented control using space vector pulse width modulation according to an embodiment; 
         FIG. 9  is a diagram illustrating field-oriented control using space vector pulse width modulation according to another embodiment; 
         FIG. 10  illustrates a dual three-phase motor system as an example application scenario; and 
         FIG. 11  illustrates a dual three-phase motor useable in the system of  FIG. 10 . 
     
    
    
     DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
     In the following, various embodiments will be discussed in detail below referring to the attached drawings. These embodiments are given by way of example only and are not to be construed as limiting. Features from different embodiments may be combined to form further embodiments. Variations, modifications and details described with respect to one of the embodiments are also applicable to other embodiments and will therefore not be described repeatedly. 
       FIG. 1  is a diagram illustrating a system according to an embodiment, including a pulse width modulation (PWM) pattern generator  10  which at least in one mode of operation employs techniques according to embodiments as disclosed herein and as will be described further below. 
     The system of  FIG. 1 , besides PWM pattern generator  10 , comprises a power source  11 , in case of a vehicle for example the battery of the vehicle, a three-phase power inverter generally labeled  110  and a motor  17 . A capacitor  111  may be coupled in parallel to power source  11 . 
     The three-phase power inverter  110  includes three half-bridges. A first half-bridge comprises a first high-side device M 1  and a first low-side device M 2 , a second half-bridge comprises a second high-side device M 3  and a second low-side device M 4 , and a third half-bridge comprises a third high-side device M 5  and a third low-side device M 6 . Each half-bridge is coupled between a first terminal of power source  11  and a second terminal of power source  11 . Each of high-side devices M 1 , M 3 , M 5 , comprises a respective high-side switch  12 A,  12 B,  12 C and a respective diode  13 A,  13 B,  13 C coupled in anti-parallel to the respective high-side switch  12 A,  12 B,  12 C. Likewise, each of low-side devices M 2 , M 4  and M 6  comprises a respective low-side switch  14 A,  14 B,  14 C and a respective diode  15 A,  15 B,  15 C coupled in anti-parallel to the respective low-side switch  14 A,  14 B,  14 C. In some embodiments, switches  12 A- 12 C and  14 A- 14 C may be implemented as transistors, for example insulated gate bipolar transistors (IGBTs), bipolar junction transistors (BJTs) or field effect transistors like metal oxide semiconductor field effect transistors (MOSFETs). Diodes  13 A- 13 C and  15 A- 15 C may be separately provided diodes or, in some cases, may be diodes part of the transistor design of the respective switch, for example body diodes. Switches  12 A- 12 C,  14 A- 14 C and diodes  13 A- 13 C,  15 A- 15 C are collectively referred to as power devices herein. Therefore, power inverter  110  in the embodiment of  FIG. 1  comprises 12 such power devices. 
     Power inverter  110  has three output nodes  112 A,  112 B,  112 C, each located between a respective pair of high-side device and low-side device, as shown in  FIG. 1 . The half-bridges and their respective output nodes are also referred to as phases U, V and W, respectively, herein, and the current flowing via the respective output node is also referred to as phase current. Motor  17  comprises three windings  18 A,  18 B,  18 C. Windings  18 A- 18 C may be stator windings, while a rotor has permanent magnets, in some embodiments. In other embodiments, windings  18 A- 18 C may be rotor windings. A first end of winding  18 A is coupled to output node  112 A, a first end of winding  18 B is coupled to output node  112 B, and a first end of third winding  18 C is coupled to output node  122 C, i.e., in operation, each of the three phase currents is provided to an associated winding  18 A- 18 C. Second ends of windings  18 A,  18 B and  18 C are coupled together. In operation, high-side switches  12 A- 12 C and low-side switches  14 A- 14 C are driven by pulse width modulated signals pwm output by PWM pattern generator  10 , causing current flow to motor  17 , which in turn causes windings  18 A- 18 C to generate magnetic fields, which generate a motor torque. The pulse width modulated signals pwm are generated based on a field-oriented control scheme using space vectors, as will be explained later in greater detail, based on a feedback signal fb indicating an angular position of the rotor of motor  17  received via a feedback path  19 , i.e., a feedback angle. Such an angular position may be measured by conventional sensors. 
     In at least one mode of operation, PWM pattern generator  10  is configured to generate signals pwm in a way that in each control period, at least four power devices take turns in bearing a full current during application of a null vector, where all three half-bridges are controlled in the same manner, as further explained later, during a control period. Such a mode of operation may be for example a mode for a low rotor speed, in particular a case where the rotor is locked, but also may be employed in other situations. A control period, as will be described later in greater detail, is a period during which a certain sequence of vectors is applied to determine the signals pwm. After the control period, as long as the angular position of the rotor is in a same sector, the sequence of vectors is repeated in a next control period. A full current is essentially a maximum current flowing through the power inverter at a given time. To be more precise, the full current is an absolute current value of the maximum phase current of the three phase currents (currents through nodes  112 A- 112 C in  FIG. 1 ) during charging motor windings or discharging of motor windings, i.e., throughout a complete control period, where the full current may be an average value in a control period or a transient value at any time of the control period. In many control schemes, at each given time, one of the power devices bears the sum of currents flowing through two other power devices. For example, in a situation as shown in  FIG. 1  where switches  12 A- 12 C are open (non-conducting between their respective load terminals) and switches  14 A- 14 C are closed (conducting between their terminals, a current may flow via diode  15 C to motor  17 , which is a sum of currents flowing from motor  17  via switches  14 A,  14 B as shown). In other phases of a control period, similar situations may occur, where a current via one of the power devices (the full current) is a sum of currents flowing via two other power devices. 
     PWM pattern generator  10  may be implemented using software, hardware, firmware or combinations thereof. For example, PWM pattern generator  10  may be implemented using one or more processors programmed by a corresponding program code, but may also be implemented using hardware like application-specific integrated circuits (ASICs) or field programmable gate arrays (FPGAs). 
       FIG. 2  is a flowchart illustrating a method according to an embodiment. The method of  FIG. 2  may be implemented in PWM pattern generator  10  of  FIG. 1 , but may also be implemented independently therefrom. In some embodiments, the method of  FIG. 2  may be implemented using a program code, which may for example be provided on a tangible storage medium, and which, when running on a processor, causes the method of  FIG. 2  to be carried out. Implementations fully or partially in hardware, for example using ASICs, FPGAs or other specific hardware, are also possible. 
     At  20  in  FIG. 2 , the method comprises detecting a low rotor speed condition or a locked rotor condition. For example, it may be detected when the rotor speed of a motor is below a predefined threshold, for example at or near zero indicating a locked rotor condition. 
     At  21 , upon detecting the low rotor speed condition or the locked rotor condition at  20 , power devices of a three-phase power inverter, for example the power devices of power inverter  110  of  FIG. 1 , are controlled such that at least four power devices take turns in carrying a full current in each control period while null vectors are applied, as explained briefly above for the system of  FIG. 1 . It should be noted that in other embodiments, detecting the low rotor speed condition at  20  may be omitted, and the control at  21  may be performed irrespective of the condition of the motor, in particular a rotor thereof. 
     Next, control techniques for power devices of a three-phase power inverter according to some embodiments, which may be used to control the power devices such that at least four power devices take turns in carrying a full current in each control period while null vectors are applied, will be described in more detail. For better understanding, first referring to  FIGS. 3-7 , field-oriented control using space vector pulse width modulation will be described in general, and the problem of hotspots in case of a locked motor condition will be explained in some detail. Following this, various non-limiting embodiments will be described. 
       FIG. 3  shows six basic active vectors {right arrow over (V)} 1  to {right arrow over (V)} 6  and six sectors 1-6 for an electric period. An electric period corresponds to a full rotation of the rotor by 360°. Each of the active vectors {right arrow over (V)} 1  to {right arrow over (V)} 6  are associated with a respective angle. For example, the angle of {right arrow over (V)} 1  is 0°, the angle of {right arrow over (V)} 2  is 60°, the angle of {right arrow over (V)} 3  is 120°, the angle of {right arrow over (V)} 4  is 180°, the angle {right arrow over (V)} 5  is 240° and the angle of {right arrow over (V)} 6  is 300°. In addition, two so-called null vectors {right arrow over (V)} 0 =[000] and {right arrow over (V)} 7 =[111] are used. The three digits of the vector indicate the control of the high-side switches of a three-phase power inverter (for example high-side switches  12 A,  12 B,  12 C in  FIG. 1 ), a “1” indicating a closed switch and a “0” indicating an open switch. The corresponding low-side switch is controlled in an inverse manner to the respective high-side switch, i.e., when the high-side switch of a half-bridge is closed, the low-side switch is open and vice versa. With null vectors, therefore all three half-bridges are controlled in the same manner. 
     The control within a control period depends on the sensed angle of the rotor, also referred to as electrical degree. When for example the sensed angle is 240°, this corresponds to vector {right arrow over (V)} 5 =[000]. This means that for the first half-bridge (phase U), the high-side power switch ( 12 A) is open and the low-side switch ( 14 A) is closed, for the second half-bridge (phase V) also the high-side power switch ( 12 B in  FIG. 1 ) is open and the low-side power switch ( 14 B in  FIG. 1 ) is closed, and for the third half-bridge (phase W) the high-side power switch ( 12 C in  FIG. 1 ) is closed and the low-side power switch ( 14 C in  FIG. 1 ) is open. 
     When an instant angle does not correspond to any of the basic active vectors, for example corresponds to vector {right arrow over (V)} ref  of  FIG. 3 , the vectors delimiting the sector in which the instant angle is used for control. For example, {right arrow over (V)} ref  is in sector 1, so the vectors {right arrow over (V)} 1  to {right arrow over (V)} 2  are used for control according to a pulse width modulated scheme, together with the null vectors {right arrow over (V)} 0  and {right arrow over (V)} 7 . For example, in a given sector k (k=1, 3, 5; i.e., odd sector number), the control scheme may be according to {right arrow over (V)} 0 -&gt;{right arrow over (V)} k -&gt;{right arrow over (V)} k+1 -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} k+1 -&gt;{right arrow over (V)} k -&gt;{right arrow over (V)} 0 . An example for the vector {right arrow over (V)} ref  in sector 1 is shown in  FIG. 4 . Here, the control transitions from {right arrow over (V)} 0 =[000] to {right arrow over (V)} 1 =[100] to {right arrow over (V)} 2 =[110] etc. Signals “pwm phase U”, “pwm phase V” and “pwm phase W” show the control signals for the three phases U, V, W of the three-phase power inverter for example as shown in  FIG. 1 , where a high signal indicates a closed high-side switch and open low-side switch, while a low signal indicates an open high-side switch and closed low-side switch, and also corresponds to a voltage at the respective output node (high or low), e.g., output nodes  112 A- 112 C of  FIG. 1 . For sectors k=2, 4, 6; i.e., even sector number, in the above sequence {right arrow over (V)} k  and {right arrow over (V)} k+1  are exchanged. 
     To give more details,  FIG. 4  shows control over a control period Ts. As used herein, Ts will be used both to refer to the control period and to the time duration thereof. Times T0, Tk and Tk+1 indicate the durations during which the respective vectors are applied, as shown in  FIG. 1 . For example, in  FIG. 4  first the null vector {right arrow over (V)} 0  is applied for T0/2, then {right arrow over (V)} 1  is applied for a time duration Tk, then {right arrow over (V)} 2  is applied for a time duration Tk+1 etc. Tk and Tk+1 are calculated according to the angle between vector {right arrow over (V)} ref , i.e., the current vector, and {right arrow over (V)} 1  ({right arrow over (V)} k ) in general) and a target voltage amplitude of the vector {right arrow over (V)} ref . For example, the closer to {right arrow over (V)} 1  {right arrow over (V)} ref  is, the longer Tk is compared to Tk+1. To is then equal to Ts/2−Tk−Tk+1. 
     When the angle of the vector {right arrow over (V)} ref  corresponds to one of the six vectors {right arrow over (V)} 1  to {right arrow over (V)} 6 , a similar control scheme as shown in  FIG. 4  is used, but the times Tk and Tk+1 are merged to a single time Tkk where the corresponding basic vector is applied. 
     The corresponding control frequency Fs=1/Ts may be for example 8 kHz for middle and high motor speeds, may be changed to 4 kHz for low motor speeds, and changed to 2 kHz for very low motor speeds including a locked rotor case with high torque output. In other words, Ts may be changed depending on predefined thresholds. It should be noted that the control scheme illustrated in  FIG. 4  may also be used in some embodiments in some other modes of operation, for example at higher rotor speeds, when no locked rotor or very low rotor speed is detected. 
     Next, a case where the rotor is locked will be explained in more detail.  FIG. 5  shows a reference example where the motor is locked at an angle of 240° corresponding to active vector {right arrow over (V)} 5 =[001]. For ease of explanation,  FIG. 5  will be described referring to  FIG. 1 . A double arrow  50  denotes the control period Ts, which is divided in time slots I-V. A curve  51  shows the control signal for phase U, a curve  52  shows the control signal for phase V and a curve  53  shows a control signal for phase W. During times Tkk, the vector {right arrow over (V)} 5  is applied. A curve  54  shows the current of phase W, including the changing current (rising part of curve  54 ) caused by applying the control vector {right arrow over (V)} 5  during the time period Tkk, where high-side switch  12 C is closed to generate current flow. Numeral  55  denotes the average current through the high-side switch of phase W (switch  12 C of  FIG. 1 ), numeral  56  denotes the average current through low-side switch  14 A, numeral  57  denotes the current flow through low-side switch  14 B, numeral  58  denotes the current through diode  13 A, numeral  59  denotes the current through diode  13 B, and numeral  510  denotes the current through diode  15 C of  FIG. 1 . Thicker bars illustrate the aforementioned full current, while thinner bars illustrate a partial current. At  511  in  FIG. 5 , current flow through the device system of  FIG. 1  is shown for each of phases I to V. For example, during phase II the full current flows through high-side switch  12 C which is closed, being a sum of currents through low-side switch  14 A and low-side switch  14 B. Likewise, for example during phase V, full current flows through diode  15 C, which is a sum of currents through low-side switches  14 A,  14 B, as can be seen by the diagrams at  511 . As mentioned, the waveforms  51 ,  52  and  53  are also indicative of the output voltage of the nodes  112 A,  112 B and  112 C, which are at a positive potential (high signal in  FIG. 5 ) when the respective high-side switch is closed, and at low potential when the respective low-side switch is closed, corresponding to the function of the half-bridges. It should also be noted that the waveforms are shown in an ideal manner, whereas in actual implementations, for example edges may have other forms than the vertical edges shown in the Figures. 
     In  FIG. 5 , Tk and Tk+1 are combined as one timeslot Tkk as explained above because the waveform of the pulse width modulation signal of phase U is completely the same as the waveform of the signal of phase V at the angle of 240°. In other words, rising and falling edges of the pulse width modulation signal of phase U ( 51  in  FIG. 5 ) are at the same points in time as rising and falling edges of phase V (signal  52  of  FIG. 5 ). This phenomenon that two of the three pulse width modulated signals of phases U, V and W are the same applies to all cases where an instant angle position of the motor coincides with one of the vectors {right arrow over (V)} 1  to {right arrow over (V)} 6 , i.e., with one of the basic active vectors. 
     Generally, when the rotor is locked, (locked rotor torque case) high current flows through the motor winding for providing a locked rotor torque, and charging time of the motor windings through time period Tkk is very short, as there is no electromagnetic force voltage on the winding due to no spinning of the rotor. For example, the two periods with lengths Tkk may have a duration of about 10% of the control period Ts as shown in  FIG. 5  or less. The exact length of Tkk changes according to different input parameters like battery voltage, resistance and inductance of the stator of the motor, required current to provide the locked rotor torque. 
     The following more detailed analysis of the situation of  FIG. 5  starts at time t1 with time slot II. Here, vector {right arrow over (V)} 5  is applied. As mentioned, and as indicated at  55 , the motor winding connected to phase W receives all current from the inverter. At t1, the current source (for example current source  11  of  FIG. 1 ) outputs energy to charge the motor windings via the closed high-side switch  12 C, flowing back via the closed low-side switches  14 A and  14 B. Therefore, high-side switch  12 C carries the full current, while low-side switches  14 A,  14 B each carry about half the full current. 
     Between t3 and t5 in time slot III, the null vector {right arrow over (V)} 7 =[111] is applied for a duration of T0. Here, low-side switches  14 A,  14 B are opened, and high-side switches  12 A,  12 B are closed. High-side switch  12 C remains closed, and low-side switch  14 C remains open. A freewheeling current due to stored energy in the motor winding flows as illustrated at  511  for time slot III, where high-side switch  12 C carries the full current, and diodes  13 A,  13 B carry about half the current. 
     Between t5 and t7 during time slot IV, the motor windings are charged again from the current source, as was explained for time slot II above. 
     When in time slot V the null vector {right arrow over (V)} 0 =[000] is applied for T0/2 and then again for T0/2 in a next time slot I of a next control period, i.e., applied altogether for a duration T0. All low-side switches  14 A,  14 B,  14 C are closed, and high-side switches  12 A,  12 B,  12 C are opened. A freewheeling current due to stored energy in the motor windings flows via low-side switches  14 A,  14 B and diode  14 C as shown at  511  for time slots V, I and as also shown in  FIG. 1 . In this case, diode  15 C carries the full current, and low-side switches  14 A,  14 B each carry about half the current. It should be noted that in implementations where an IGBT is used as switch, the switch  14 C is reversed biased, so essentially all the current flows via the diode. In other switch implementations like MOSFETs, in principle current could also flow via the closed switch  14 C, but diode  15 C in usual implementations carries at least most of the current due to lower resistance. In the next control period the same action repeats. As the motor is in a rotor-locked condition or in a condition with very slow rotational speed, also the motor angle does not progress or does not progress fast to a next sector of the field-oriented control scheme (see  FIG. 6 ), such that the control period illustrated with respect to  FIG. 5  may be repeated many times. 
     As can be seen in  FIG. 5 , not all twelve power devices are active (carry current) at the locked-rotor torque case, but only six power devices are involved. Moreover, among the six power devices involved, the average current when conducting is not the same. In the example of  FIG. 5 , switch  12 C and diode  15 C carry the full current, whereas other power devices involved carry only about half this full current. Moreover, for the power devices involved, the time during which they are conducting current is not the same. For example, high-side switch  12 C as seen in  FIG. 5  carries current for T0+2*Tkk, whereas diode  15 C carries the full current for a duration of T0. This means that the duty cycle for switch  12 C may be about 55%, assuming that 2*Tkk is about 10% of Ts, and the duty cycle of diode  15 C is about 45%. 
     If assuming that the voltage across switch  12 C is about the same as the voltage drop across diode  15 C when carrying the full current, conduction power loss of diode  15 C due to the different duty cycles is about 82% (45/55) of the power loss in high-side switch  12 C. Therefore, high-side switch  12 C may become hottest (hottest hotspot), and diode  15 C is the second hottest hotspot. Other power devices involved, as they carry only about half the full current, are less critical. 
     In some conventional implementations to reduce problems with hotspots, balancing power loss between the two hottest devices (in the example of  FIG. 5  high-side switch  12 C and diode  15 C) is performed. For example, in  FIG. 5  to achieve this, the duration T0 of time slot III where the vector {right arrow over (V)} 7  is applied is reduced, and the duration of the two time slots I, V T0/2 where the vector {right arrow over (V)} 0  is applied is increased accordingly. However, as the differences in duty cycles for these devices are not very high, the effect is limited. In particular, in the numerical example given above, in this case the duty cycle for high-side switch  12 C would be reduced from 55% to 50%, which is a comparatively low reduction of power loss. Furthermore, this approach is only feasible if the instant angular position of the rotor corresponds to one of basic active vectors {right arrow over (V)} 1  to {right arrow over (V)} 6 . 
     Before turning to the techniques for reducing hotspots according to various embodiments, with reference to  FIG. 6  the more general case where the instant angle of the rotor is in any of sectors 1-6 of  FIG. 3  without coinciding with one of the vectors {right arrow over (V)} 1  to {right arrow over (V)} 6  will be discussed referring to  FIG. 6 . 
       FIG. 6  shows an example where the angle is in sector 4 of  FIG. 3 , with vector {right arrow over (V)} k ={right arrow over (V)} 4 =[011] and vector {right arrow over (V)} k+1 ={right arrow over (V)} 5 =[001]. In  FIG. 6 , numeral  50  again denotes the control period, a curve  61  shows the control for phase U (similar to curve  51  of  FIG. 5 ), a curve  62  shows the control for phase V (similar to curve  52  of  FIG. 5 ), and a curve  63  shows the control for phase W (similar to curve  53  of  FIG. 5 ). A curve  64  shows the current of phase W and/or U, corresponding to a current flowing through  112 C and/or  112 B of  FIG. 1 . A main difference to  FIG. 5  is that each of the time slots with duration Tkk where vector {right arrow over (V)} 5  is applied, is replaced by two time slots with durations Tk+1 and Tk, where the vectors {right arrow over (V)} 5  and {right arrow over (V)} 4  are applied (time slots II, III and V, VI of  FIG. 6 ). Numeral  62  denotes the average current through high-side switch  12 C (similar to  55  of  FIG. 5 ), numeral  66  denotes the average current through low-side switch  14 A (similar to  56  of  FIG. 5 ), numeral  67  denotes the average current through low-side switch  14 B (similar to  57  of  FIG. 5 ), numeral  68  denotes the average current through diode  13 A (similar to  58  of  FIG. 5 ), and numeral  610  denotes the average current through diode  15 C is shown (similar to  510  of  FIG. 5 ). The vector {right arrow over (V)} 5  is applied in time slots II, VI, and the vector {right arrow over (V)} 4  is applied in time slots III, V. 
     Similar to  FIG. 5 , also in the situation of  FIG. 6  the charging time (time slots II, III, V, VI) are a comparatively small part of a control period, in the example shown in  FIG. 6  about 10% of Ts, as in  FIG. 5 . Furthermore, for the example of  FIG. 6  it is assumed that Tk=Tk+1. This is for example exactly the case if the vector {right arrow over (V)} ref  is exactly between {right arrow over (V)} 4  and {right arrow over (V)} 5 . For other positions, the relationship may vary. Furthermore, the proportion of the total charging time (2Tk+2Tk+1) in a control period Ts depends on input parameters like supply voltage, resistance and inductance of the stator of the motor (for example windings  18 A- 18 C of  FIG. 1 ) or current needed through provide the locked rotor torque. 
     Explanation of the control scheme of  FIG. 6  starts in time slot II, where the vector {right arrow over (V)} 5  is applied for a time Tk+1. Again, for convenience reference will be made to the system of  FIG. 1  for ease of explanation. In phase II, current source  11  outputs energy to charge the motor windings via closed high-side switch  12 C and low-side switches  14 A,  14 B, where high-side switch  12 C carries the full current ( 65  in  FIG. 6 ), whereas low-side switches  14 A,  14 B each carry about the half current. 
     During time slot III, current source  11  continues to output energy to charge the motor windings, in this case via high-side switches,  12 B,  12 C which are closed and low-side switch  14 A which is closed. In this case ( 66  in  FIG. 6 ), low-side switch  14 A carries the full current, and high-side switches  12 B,  12 C ( 65 ,  67  in  FIG. 6 ) each carry about half the full current. 
     During time slot IV, the null vector {right arrow over (V)} 7 =[111] is applied. High-side switches  12 A- 12 C are closed and low-side switches  14 A- 14 C are open. In this case, a freewheeling current due to stored energy in the motor windings flows as shown for time slot IV at  611  of  FIG. 4 . Diode  13 A carries the full current ( 69  in  FIG. 6 ), whereas high-side switches  12 B,  12 C each carry about half the full current ( 65 ,  68  in  FIG. 6 ). 
     In time slot V, the situation is essentially the same as in time slot III, where also the vector {right arrow over (V)} 4  is applied. As in time slot III, low-side switch  14 A carries the full current, whereas high-side switches  12 B,  12 C each carry about half the current. 
     In time slot VI, the charging continues, where the situation essentially corresponds to the situation in time slot II, where also the vector {right arrow over (V)} 5 =[001] is applied. As in time slot II, high-side switch  12 C carries the full current and low-side switches  14 A,  14 B each carry about half the full current. 
     In time slot VII and a next time slot I, the null vector {right arrow over (V)} 0 =[000] is applied for a time T0 (T0/2 in time slot VII and T0/2 in time slot I). The freewheeling current from the motor windings flows via low-side switches  14 A,  14 B and diode  15 C as shown at  611  for time slots VII, I. Diode  15 C carries the full current, and low-side switches  14 A,  14 B each carry about half the full current. 
     In the next control period Ts, the same action repeats as long as the rotor is locked. The following features and properties may be deduced from the example of  FIG. 6 . 
     First of all, similar to  FIG. 5 , not all twelve power devices of the power inverter carry current during a locked rotor case, but there are only six power devices involved. Moreover, among these six power devices, the average current of each one conducting is not the same. In the example of  FIG. 6 , only high-side switch  12 C, low-side switch  14 A, diode  13 A and diode  15 C carry the full current, whereas other power devices only carry about half the full current. 
     However, among these four power devices carrying the full current, the times during which they carry the full current differs significantly. The time during which high-side switch  12 C and low-side switch  14 A carry the full current during a control period Ts is very short (2Tk+1 and 2Tk, respectively), which corresponds to a duty cycle of about 5%. The time during which diode  13 A and diode  15 C carry the full current is significantly longer, each for a period T0 corresponding to a duty cycle of 45%. 
     If similar as in the example of  FIG. 5  it is assumed that the voltage drop is about the same for all twelve power devices, the conduction power loss of each power device is proportional to the duty cycle and the current carried during the current cycle. Therefore, in the example of  FIG. 6  diodes  13 A and  15 C have by far the highest conduction power losses, whereas the power losses for the other four power devices involved is much lower. Therefore, these power devices create the most heat and form hotspots. Moreover, as their duty cycle is at least approximately the same, a balancing between the duty cycles between these two power devices, as explained as a conventional method for the situation in  FIG. 5 , is hardly possible. 
     For the other five sectors ( FIG. 6  shows an example for sector 4 as mentioned), a similar analysis can be performed, and in each case two of the diodes have the highest power losses. An overview is given in  FIG. 7  which essentially reproduces  FIG. 3  and additionally states which diodes have the highest power losses for each sector, each conducting the full current via a period T0. 
     For the above explanations, it can also be deduced that the reason why the conduction power loss at a locked rotor torque case is higher than in case of a low rotor speed with the same torque. To explain this, diode  15 C is used as an example. Diode  15 C is one of the hotspot devices in sectors 4 and 5, but not in any of the other sectors. If the motor is rotating (even when it is slow), the target vector position ({right arrow over (V)} ref  of  FIG. 3 ) also moves in the vector map through sectors 1-6. Therefore, in this case diode  15 C is a hotspot device only in two of the six sectors, which gives an overall duty cycle of about 0.15 in an electric period (⅓*0.45)TE, i.e., one revolution of the motor, which is much lower than the duty cycle of 45% at the locked rotor torque case. Nevertheless, techniques discussed below may, e.g., also be applied to a case where the rotor is spinning with low speeds or in other situations. 
     In embodiments, to reduce power losses in at least one mode of operation, e.g., in a locked rotor case as already briefly mentioned with respect to  FIGS. 1 and 2 , in embodiments a three-phase power inverter is controlled by a PWM pattern generator like PWM pattern generator  10  of  FIG. 1  such that at least four power devices of the three-phase power inverter take turn in conducting the full current while a null vector ({right arrow over (V)} 0 =[000] or {right arrow over (V)} 7 =[111] in the examples above) is applied. In other words, at least four power devices of the three-phase power inverter take turn in conducting a full current during a comparatively large part of the control period, for example during at least 60% of the control period or more, like during at least 80 &amp;% or at least 90% of the control period Ts. In this way, conduction power losses in individual power devices may be reduced in some embodiments. 
     Control schemes according to embodiments discussed in the following are based on the two null vectors {right arrow over (V)} 0  and {right arrow over (V)} 7  and on the two basic active vectors delimiting a sector in which the angle corresponding to an instant rotor position is located (for example {right arrow over (V)} 1  and {right arrow over (V)} 2  when the vector {right arrow over (V)} ref  is in sector 1, etc.). Various approaches to implement such a control scheme will be discussed below: 
     Approach 1: For a first approach of a control scheme according to some embodiments, four different combinations of two vectors are defined, wherein in each combination one of the basic active vectors delimiting a respective sector is followed by one of the null vectors. As before, the two basic active vectors delimiting a sector will be named {right arrow over (V)} k  and {right arrow over (V)} k+1 , and the null vectors are {right arrow over (V)} 0  and {right arrow over (V)} 7 . The four vector combinations are then {right arrow over (V)} k -&gt;{right arrow over (V)} 0  (i.e., transition from {right arrow over (V)} k  to {right arrow over (V)} 0 ), {right arrow over (V)} k -&gt;{right arrow over (V)} 7 , {right arrow over (V)} k+1 -&gt;{right arrow over (V)} 0  and {right arrow over (V)} k+1 -&gt;{right arrow over (V)} 7 . No vector is inserted between the vectors of the combination. In the first approach, in each control period Ts all four of these four combinations of two vectors are applied at least once. 
     In particular, in some embodiments the four combinations may be applied in sequences, without additional control vectors, wherein the order in which the four vector combinations are applied may be varied. 
     An example for this approach 1 will be discussed later referring to  FIG. 9 . 
     Approach 2: Also in approach 2, the two basic active vectors {right arrow over (V)} k  and {right arrow over (V)} k+1  are used together with the two null vectors {right arrow over (V)} 0  and {right arrow over (V)} 7 . For a control sequence, two combinations of three vectors are defined, wherein one of the combination comprises one of active vectors, for example {right arrow over (V)} k , followed by the two null vectors ({right arrow over (V)} 0  and {right arrow over (V)} 7 , in any order), and the other combination of three vectors comprises the respective other basic active vector, for example {right arrow over (V)} k+1 , followed by the two different null vectors in any order. For example, the combinations may be {right arrow over (V)} k -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7  and {right arrow over (V)} k+1 -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 . Instead of the order {right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 , or also the order {right arrow over (V)} 7 -&gt;{right arrow over (V)} 0  may be used in one or both of the sequences. Both three vector combinations are then applied in a control sequence. In some embodiments, no further vectors are used. In other embodiments, additional vectors may be inserted between the two sequences, but not within the sequences. 
     It should be noted that this approach 2 is related to approach 1 in so far as each vector combination in some sense “combines” two of the combinations of two vectors of approach 1. For example, {right arrow over (V)} k -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7  may be seen as a combination of {right arrow over (V)} k -&gt;{right arrow over (V)} 0  and {right arrow over (V)} k -&gt;{right arrow over (V)} 7 . A specific example for this approach 2 will be later explained referring to  FIG. 8 . 
     Approach 3: Approach 3 is a mix of the approaches 1 and 2. Here, one of the combinations of three vectors of approach 2 is used, together with two of the combinations of two vectors of approach 1, in each control period. In some embodiments, the two combinations of two vectors used are those of the active vector not used in the combination of three vectors. For example, as combination of three vectors {right arrow over (V)} k -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0  may be used, and in addition the two combinations of two vectors {right arrow over (V)} k+1 -&gt;{right arrow over (V)} 0  and {right arrow over (V)} k+1 -&gt;{right arrow over (V)} 7  may be used. 
     After these explanations of the different approaches, specific examples for these approaches will be discussed referring to  FIGS. 8 and 9 . The way of representations in the diagrams of  FIGS. 8 and 9 , for ease of comparison and for better understanding, corresponds to the way the reference examples were discussed in  FIGS. 5 and 6 . 
       FIG. 8  illustrates a control scheme based on approach 2 above, using two combinations of three vectors, in this case with additional vectors inserted between the combinations. Numeral  50  again denotes the control period Ts. Each control period in this case may be divided into eight time slots labeled I-VIII, in which different control vectors are applied successively. Curves  81 ,  82  and  83  show the control of phases U, V, W similar to curves  51 - 53  of  FIG. 5  and curves  61 - 63  of  FIG. 6  and may therefore also illustrate a voltage at nodes  112 A,  112 B and  112 C of  FIG. 1 , respectively. Furthermore, similar to curves  54  and  64 , curve  84  shows a current for phase W and/or U, e.g., a current flowing via output node  112 C of  FIG. 1 . 
       FIGS. 8 and 9  each illustrate a case where an angular position of the rotor is in sector 4, i.e., {right arrow over (V)} ref  is in sector 4, such that {right arrow over (V)} 4  and {right arrow over (V)} 5  are the basic active vectors delimiting the sector. Numeral  85  denotes the average current through high-side switch  12 C, numeral  86  denotes the average current through low-side switch  14 A, numeral  87  denotes the average current through low-side switch  14 B, numeral  88  denotes the average current through high-side switch  12 B, numeral  89  denotes the average current through diode  13 A, numeral  810  denotes the average current through diode  15 C, numeral  811  denotes the average current through diode  15 B and numeral  812  denotes the average current through diode  13 B. As before, thicker bars indicate full current, whereas thinner bars indicate about half the full current flowing. 
     At  813 , essentially the power converter and motor of  FIG. 1  are reproduced, showing the current flow in each phase. 
     In  FIG. 8 , similar to  FIGS. 5 and 6 , it is assumed that the total charging time where energy flows from battery current source  11  to the motor is about 10% of the control period Ts, corresponding to time slots II, III, VI and VII in  FIG. 8 . Furthermore, as for  FIG. 6 , for the subsequent analysis it is assumed that Tk+1 and Tk are equal. The real value, as explained for  FIG. 6 , may be depending on parameters like instant angle, battery voltage, resistance and inductance of motor stator and current needed to provide the locked rotor torque. 
     The following analysis starts in time slot  2 . Here, the vector {right arrow over (V)} 5 =[001] is applied. Current source  11  outputs power to charge windings  18 A,  18 C of motor  17  via high-side switch  12 C, low-side switch  14 A and low-side switch  14 B, where high-side switch  12 C carries the full current and low-side switches  14 A,  14 B each carry about half the current. 
     In time slot III, vector {right arrow over (V)} 4 =[011] is applied, continuing the charging. Here, current source  11  continues to output energy to charge the motor windings via high-side switches  12 B,  12 C and low-side switch  14 A. Low-side switch  14 A carries the full current, and high-side switches  12 B,  12 C carry about half the full current. 
     In time slot IV, the null vector {right arrow over (V)} 7 =[111] is applied for T0/2. Compared to time slot III, low-side switch  14 A is opened and high-side switch  12 A is closed, so that all high-side switches are closed. A freewheeling current flows as shown at  813  for phase IV via diode  13 A and high-side switches  12 B,  12 C. Diode  13 A carries the full current, whereas high-side switches  12 B,  12 C each carry about half the current. 
     During time slot V, the null vector {right arrow over (V)} 0 =[000] is applied, opening all high-side switches  12 A to  12 C and closing all low-side switches  14 A- 14 C. Freewheeling current flows as shown at  813  for phase V. Low-side switch  14 A carries the full current, while diodes  15 B and  15 C each carry about half the full current. 
     After this, in time slots VI and VII, the motor is charged again by application of vector {right arrow over (V)} 4  followed by vector {right arrow over (V)} 5 . In time slot VI, similar to time slot III, low-side switch  14 A carries the full current, while high-side switches  12 B,  12 C each carry about half the full current. During time slot VII, similar to time slot II, high-side switch  12 C carries the full current, while low-side switches  14 A,  14 B each carry about half the current. 
     In time slot VIII, again the null vector {right arrow over (V)} 0 =[000] is applied. In this case, the freewheeling current flows via low-side switches  14 A and  14 B as well as diode  15 C. Diode  15 C carries the full current, while low-side switches  14 A,  14 B each carry about half the full current. Following this, in time slot I of a next control period Ts, the null vector {right arrow over (V)} 7 =[111] is applied, closing all high-side switches and opening all low-side switches. Here, high-side switch  12 C carries about the full current, while diodes  13 A,  13 B each carry about half the full current. 
     Then, the above described sequence is repeated. As already mentioned,  FIG. 8  shows an example for approach 2 mentioned above. The first combination of three vectors is applied in time slots III, IV and V as {right arrow over (V)} 4 -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0 , and the other combination of three vectors is applied in time slots VII, VIII and the next time slot I, as {right arrow over (V)} 5 -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 . Therebetween, in time slots II and VI, the respective other active vector delimiting the instant sector is applied. 
     In the example below, still not all twelve power devices carry current in the locked rotor torque case, but there are eight power devices involved. Of these eight power devices, there are four power devices carrying the full current, namely high-side switch  12 C, low-side switch  14 A, diode  15 A and diode  15 C. Each of these power devices, in contrast for example to  FIG. 6 , carry the full current while a null vector is applied, leading to a more even distribution of duty cycles among these four power devices. Using the numerical examples given above, the duty cycle of high-side switch  12 C and low-side switch  14 A for carrying the full current are each 27.5%, and the duty cycles carrying the full current for diodes  13 A and  15 C are each 22.5% of the control period. Therefore, these four power devices take turns in bearing the full current, and the maximum duty cycle a device bears the full current is reduced compared for example to  FIG. 6 . It should be taken into account that each of the four power devices also bears about half the full current for some time, which also contributes to some power losses. 
     To analyze more precisely and taking into account that these devices also bear half the full current during some time slots, when U is the voltage drop across each power device, I is the average value of the full current and it is assumed that the voltage drop across all 12 power devices is the same, the power losses P for the aforementioned devices may be calculated as follows: 
         P (high-side switch 12 C )=( U*I* 22.5%* Ts+U*I* 2.5%* Ts+U*I* 2.5%* Ts+U* 0.5* I* 22.5%* Ts+U* 0.5* I* 2.5%* Ts+U* 0.5* I* 2.5%* Ts )/ Ts= 41.25%* U*I.    
     The power loss for low-side switch  14 A, P (low-side switch  14 A) is the same as P (high-side switch  12 C) and therefore also 41.25%*U*I. 
     The power loss for diode  13 A and for diode  15 C each is: 
         P (diode 13 A )= P (diode 15 C )=( U*I* 22.5%* Ts+U* 0.5* I* 22.5%* Ts )/ Ts= 33.75%* U*I.    
     The above calculations are for a charging time proportion of 10%, i.e., (2*Tk+2*Tkk)=0.1*Ts. 
     The value for the power losses changes with parameters. As an example, below the power losses are calculated for a total charging time making up 5% of the control period Ts (2*Tk+2*Tkk=0.05*Ts), and 5% ripple of the full current. This is a realistic scenario for many applications, as for many applications in the locked rotor torque case the charging time is less than 10% and may be about 5% of the control period. For example, the inductance of each of the three motor windings  18 A to  18 C may be about 500 μH. The control frequency 1/Ts in such a case may be 2 kHz. This means the control period Ts is about 500 μs. In such a situation, the charging time from 95% to 105% of the average full current may be about 15 μs, which is 3% of Ts. In addition, an average value for carrying the full current via the switches is 2.5% less than the average value of the full current in Ts. The average value for carrying the full current via one of the diodes is 2.5% higher than the average value of the full current in Ts. For example, during time slot IV, the full current via diode  13 A may be 2.5% higher than the average full current during Ts, and during time slot V the average value for the full current via low-side switch  14 A may be 2.5% lower than the average full current over the complete control period Ts. This gives an overall variation of the full current of 5%, being the above-mentioned ripples. This leads to the following results for the power losses: 
         P (high-side switch 12 C )= P (low-side switch 14 A )=( U* 0.975* I* 23.75%* Ts+U* 0.975* I* 1.25%* Ts+U* 0.975* I* 1.25%* Ts+U* 0.5* I* 23.75%* Ts+U* 0.5* I* 1.25%* Ts+U* 0.5* I* 1.25%* Ts )/ Ts= 38.72%* U*I    
         P (diode 13 A )= P (diode 15 C )=( U* 1.025* I* 23.75%* Ts+U* 0.5* I* 23.75%* Ts )/ Ts= 36.22%* U*I.    
     Therefore, in this perhaps more realistic scenario the power losses of the four power devices are more similar to each other than in the above-captioned case of 10%. As the charging time in realistic situations is more likely to be of the order of 5% than of the order of 10%, this means that usually a greater balance between the power devices than for a charging time of 10% Ts may be obtained. Furthermore, by distributing the full current and associated power losses over the four power devices in particular during times when null vectors are applied, which make up a higher proportion of Ts than the times where active vectors (charging time) are applied, power losses in individual devices may be reduced compared to the reference examples of  FIGS. 5 and 6 , therefore reducing formation of hotspots. This in some embodiments may relax the requirements for designing the power devices, which in some cases may help to save costs. 
       FIG. 9  illustrates an example for the approach 1 mentioned above, and is given with a diagram similar to the diagrams of  FIGS. 5, 6 and 8 . Numeral  50  again denotes the control period, which in this case may have a duration Ts twice the duration Ts in  FIG. 8 , as in this case a lower control frequency Fs is sufficient as will be explained below. Each control period Ts may be divided into eight time slots I to VIII. 
     In particular, when the length of the control period is doubled in  FIG. 9  compared to  FIG. 8 , the time T0 also doubles, such that the discharging periods in  FIGS. 9 and 8  have the same length. Reducing the control frequency more depending on the implementation in each case may lead to torque shape and interrupted torque, as the current ripple may increase with shorter discharge periods. 
     In  FIG. 9 , curves  91  to  93  shows the control signals for phases U, V and W corresponding to voltages at the output nodes  112 A to  112 C, as was explained for the respective curves  51  to  53  of  FIG. 5, 61 to 63  of  FIGS. 6 and 81 to 83  of  FIG. 8 . A curve  94  shows the transient and average current of phase W and, where applicable, also for phase U. Numeral  95  denotes the average current through high-side switch  12 C, numeral  96  denotes the average current through low-side switch  14 A, numeral  97  denotes the average current through low-side switch  14 B, numeral  98  denotes the average current through high-side switch  12 B, numeral  99  denotes the average current through diode  13 A, numeral  910  denotes the average current through diode  15 C, numeral  911  denotes the average current through diode  15 B and numeral  912  denotes the average current through diode  13 B. Thick bars denote the full current flowing, and thinner bars denote half the full current flowing. At  913 , current flow for the varying phases is shown. 
     Time slots I to VIII contain the four combinations of two vectors mentioned for approach 1 in sequence. In particular, in time slots I and II, {right arrow over (V)} 5 -&gt;{right arrow over (V)} 0  is applied, in time slots III and IV {right arrow over (V)} 5 -&gt;{right arrow over (V)} 7  is applied, in time slots V and VI {right arrow over (V)} 4 -&gt;{right arrow over (V)} 7  is applied, and in phases VII and VIII {right arrow over (V)} 4 -&gt;{right arrow over (V)} 0  is applied. 
     As can be seen by curve  94 , compared to for example  FIG. 8  in each control period Ts there are four charging times (during applying an active vector) and four discharging times (while applying the following null vector). Therefore, compared to  FIG. 8 , for application of the control scheme of  FIG. 9  in some embodiments the control period Ts may have twice the length than the control period of  FIG. 8 , corresponding to half the control frequency Fs. For example, when the control frequency Fs=1/Ts is 2 kHz in  FIG. 8 , it may be 1 kHz in  FIG. 9 . 
     Furthermore, as can be seen from the thick bars in  FIG. 9 , again four of a total of eight power devices carrying current carry the full current, the same power devices as in  FIG. 8 , namely high-side switch  12 C, low-side switch  14 A, diode  13 A and diode  15 C. 
     Taking a charging time proportion 5% and the control period Ts with twice the length compared to  FIG. 9 , the conduction losses in the case of  FIG. 9  are, calculated in the same manner as above: 
         P (high-side switch 12 C )= P (low-side switch 14 A )=38.4375%* U*I    
         P (diode 13 A )= P (diode 15 C )=36.5625%* U*I.    
     The following table summarizes the above calculated conduction power losses and compares them to the conventional case of  FIG. 6 : 
     
       
         
           
               
               
               
               
             
               
                   
                 TABLE 1 
               
               
                   
                   
               
               
                   
                 FIG. 8 
                 FIG. 8 
                 FIG. 9 
               
               
                   
                 conduction 
                 conduction 
                 conduction 
               
               
                   
                 power loss 
                 power loss 
                 power loss 
               
               
                   
                 (10% charg- 
                 (5% charg- 
                 (5% charg- 
               
               
                   
                 ing time) 
                 ing time) 
                 ing time) 
               
               
                   
                 (*U*I) 
                 (*U*I) 
                 (*U*I) 
               
               
                   
                   
               
             
            
               
                   
               
            
           
           
               
               
               
               
               
            
               
                 Conventional 
                 Switches 
                   30% 
                 27.5% 
                 * 
               
               
                 PWM (FIG. 6) 
                 12C, 14A 
               
               
                   
                 Diodes 
                   45% 
                 47.5% 
                 * 
               
               
                   
                 13A, 15C 
               
               
                 PWM of 
                 Switches 
                 41.25% 
                 38.72% 
                 38.4375% 
               
               
                 embodiments 
                 12C, 14A 
               
               
                   
                 Diodes 
                 33.75% 
                 36.22% 
                 36.5625% 
               
               
                   
                 13A, 15C 
               
               
                 Conduction 
                 Hotspot 
                 8.3% 
                 18.5% 
                 19.1% 
               
               
                 power loss 
                 power 
                 [(45- 
                 [(47.5- 
                 [(47.5- 
               
               
                 improvement 
                 devices 
                 41.25)/45] 
                 38.72)/47.5] 
                 38.43)/47.5] 
               
               
                 by embodiment 
               
               
                   
               
            
           
         
       
     
     In the above table, for  FIG. 6  a control frequency of 1 kHz as was applied to  FIG. 9  is not possible, therefore here for the improvement calculation 2 kHz has been used as a control frequency in  FIG. 6 . As can be seen, conduction power loss in the hotspot devices is reduced in the embodiments by 8.3%, 18.5% and 19.1%, respectively, compared to the conventional case of  FIG. 6 . In case of  FIG. 9 , the minimum control frequency needed may be about half the minimum control frequency compared to the conventional case. It should also be noted that the improvement becomes greater when the charging time is reduced (greater improvement at 5% charging time compared to 10% charging time). 
     The conduction power losses dominate the complete power losses. Nevertheless, switching power losses also may have some impact. 
     In the examples of  FIG. 8  (approach 2), switching power losses may be a bit higher than in the conventional case of  FIG. 6 , as more switching events occur. In particular, in this case in some implementations the switching frequency of power devices may be two to three times higher than in the conventional case. Nevertheless, as conduction power losses dominate compared to switching power losses, still power may be saved. For the case of  FIG. 9  (approach 1), as the control frequency may be halved, the switching power losses are roughly the same or even slightly below the conventional case. In this respect, it should be noted that the transitions between adjacent vectors in the example of  FIG. 9  is as smooth as in the conventional sequence of  FIG. 6 . 
     It should be noted that  FIGS. 5, 6, 8 and 9  show examples for sector 4, i.e., an even sector. For odd sectors, the positions of {right arrow over (V)} k  and {right arrow over (V)} k+1  may be reversed. When nor particular order is implied, the two active vectors delimiting a sector may also be referred to as {right arrow over (V)} a  and {right arrow over (V)} b . 
     In summary by the various approaches and techniques disclosed herein, power losses when driving a three-phase power inverter to control an electric motor may be reduced. 
     In the embodiments described above, a three-phase inverter is used to control a three-phase motor. This, however, is not to be construed as limiting. For example, the FOC control as discussed above may also be applied to a dual three-phase motor controlled by two three-phase inverters. This will be briefly explained referring to  FIGS. 10 and 11 . 
       FIG. 10  shows a system comprising a dual three-phase motor  1000  controlled by a first three-phase inverter  1001 A and a second three-phase inverter  1001 B. Each of three-phase inverters  1001 A,  1001 B may be controlled according to techniques discussed above, i.e., such that at least in a mode of operation like a locked rotor conditions for each three-phase inverter  1001 A,  1001 B four power devices take turn in bearing a full current during application of null vectors. Three-phase inverters  1001 A,  1001 B are supplied by a supply voltage U dc  via a filtering capacitor  1002  in the example system of  FIG. 10 . 
     A dual three-phase motor is a motor, which includes two sets of three windings. In some implementations, the two sets are electrically isolated from each other. In other implementations, the two sets may have a common electrical node. An example for the first case is shown in  FIG. 11 . 
       FIG. 11  schematically shows a motor including a first set of windings  1101 A,  1101 B and  1101 C and a second set of windings  1102 A,  1102 A and  1102 C. The first set of windings is offset to the second set of windings by an angle, which is 30° in the example of  FIG. 11 . Windings  1101 A,  1101 B and  1101 C may be supplied by phases u 1 , v 1  and w 1  from first three-phase inverter  1001 A of  FIG. 10 , respectively, and windings  1102 A,  1102 B and  1102 C may be supplied by phases u 11 , v 11  and w 11  from first three-phase inverter  1001 A of  FIG. 10 . In  FIG. 11 , windings  1101 A,  1101 B and  1101 C are electrically coupled with each other at a node  1103 A, and windings  1102 A,  1102 B,  1102 C are electrically coupled with each other at a node  1103 B. However, the first and second set of windings are not electrically connected. 
     In other embodiments, 6-phase motors may be driven in a similar manner to the dual three-phase motor explained with reference to  FIGS. 10 and 11 , with a similar inverter arrangement as shown in  FIG. 10 , which the acts as a six-phase inverter. Here, a single 6 phase control scheme is used, which may be a combination of two control schemes as discussed above for two groups of three windings. In such a six-phase motor, the windings of the motor are electrically connected at a common node. 
     Some embodiments are defined by the following examples: 
     Example 1 
     A pulse width modulation pattern generator configured to control a three-phase power inverter; 
     wherein the three-phase power inverter comprises three half-bridges each comprising two switches and two diodes coupled in anti-parallel to the switches as power devices; 
     wherein the pulse width modulation pattern generator is configured to control the three-phase power inverter using field-oriented control via space vector pulse width modulation; 
     wherein, in at least one mode of operation, the pulse width modulation pattern generator is adapted to control the three-phase power inverter such that in each control period of the space vector pulse width modulation, at least four of the power devices of the three-phase power inverter take turns in bearing a full current during application of a null vector; 
     null vectors being vectors where all three half-bridges are controlled in a same manner; and 
     wherein a full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     Example 2 
     The pulse width modulation pattern generator of example 1; 
     wherein the pulse width modulation pattern generator is configured to control the three-phase power inverter using field-oriented control via space vector pulse width modulation based on a feedback angle and control vectors selected based on the feedback angle. 
     Example 3 
     The pulse width modulation pattern generator of example 1 or 2; 
     wherein the at least one mode of operation is; 
     a mode of operation with a locked rotor condition of a motor controlled by the three-phase power inverter; or 
     a mode of operation where a rotation speed of the motor is below a predefined threshold. 
     Example 4 
     The pulse width modulation pattern generator of any one of examples 1 to 3; and 
     wherein in the at least one mode of operation, in each control period the control is based on two active vectors delimiting a sector indicated by a feedback angle and on two different null vectors. 
     Example 5 
     The pulse width modulation pattern generator of example 4; 
     wherein the pulse width modulation pattern generator is adapted to employ, in the at least one mode of operation, in each control period; and 
     four different sequences of the two active vectors and the two null vectors, each sequence including one of the two active vectors and one of the two null vectors. 
     Example 6 
     The pulse width modulation pattern generator of example 5; 
     wherein the pulse width modulation pattern generator is adapted to control the three-phase power inverter in each control period according to a control scheme {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 0 ; and 
     where {right arrow over (V)} a , {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 7 
     The pulse width modulation pattern generator of example 4; 
     wherein the pulse width modulation pattern generator is adapted to employ, in the at least one mode of operation, in each control period; 
     a first sequence including one of the active vectors followed by two different null vectors; and 
     a second sequence including the other one of the two active vectors followed by two different null vectors. 
     Example 8 
     The pulse width modulation pattern generator of example 7; 
     wherein the first sequence is one of {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7  or {right arrow over (V)} a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0 ; 
     the second sequence is one of {right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0  or {right arrow over (V)} b -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 ; and 
     where {right arrow over (V)} a , {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 9 
     The pulse width modulation pattern generator according to example 7 or 8; 
     wherein the pulse width modulation pattern generator is adapted to employ one of the active vectors between the first sequence and the second sequence. 
     Example 10 
     The pulse width modulation pattern generator according to example 4; 
     wherein the pulse width modulation pattern generator is adapted to employ, in the at least one mode of operation, in each control period; 
     two different sequences of two vectors;| 
     each of the two different sequences including one of the two active vectors and a null vector; and 
     one sequence including one of the two active vectors followed by two different null vectors. 
     Example 11 
     The pulse width modulation pattern generator according to example 10; 
     wherein each of the two different sequences includes the one of the two active vectors followed by the null vector. 
     Example 12 
     A system, comprising: 
     the pulse width modulation pattern generator of any one of examples 1 to 11, and a three-phase power inverter coupled to the pulse width modulation pattern generator. 
     Example 13 
     The system of example 12, further comprising a motor coupled to the three-phase power inverter. 
     Example 14 
     The system of example 13, wherein the motor is a dual three phase motor, the system further comprising a further three-phase power inverter coupled to the motor and to the pulse width modulation pattern generator. 
     Example 15 
     A system, comprising: 
     a six-phase power inverter, wherein the six-phase power inverter comprises six half-bridges each comprising two switches and two diodes coupled in anti-parallel to the switches as power devices; 
     a pulse width modulation pattern generator configured to control the six-phase power inverter; 
     wherein the pulse width modulation pattern generator is configured to control the six-phase power inverter using field-oriented control via space vector pulse width modulation; 
     wherein, in at least one mode of operation, the pulse width modulation pattern generator is adapted to control the six-phase power inverter such that in each control period of the space vector pulse width modulation, for each of two groups of three half-bridges of the six half-bridges at least four of the power devices of the three-phase power inverter take turns in bearing a full current during application of a null vector; 
     null vectors being vectors where all three half-bridges are controlled in a same manner; and 
     wherein a full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     Example 16 
     A method for controlling a three-phase power inverter; 
     the three-phase power inverter comprising three half-bridges each comprising two switches and two diodes coupled in anti-parallel to the switches as power devices; 
     the method comprising: 
     using field-oriented control via space vector pulse width modulation; and 
     in at least one mode of operation, controlling the three-phase power inverter such that in each control period of the space vector pulse width modulation four of the power devices take turns in bearing a full current during application of a null vector; 
     null vectors being vectors where all three half-bridges are controlled in the same manner; and 
     wherein a full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     Example 17 
     The method of example 16, wherein the using is based on a feedback angle and control vectors selected based on the feedback angle. 
     Example 18 
     The method of example 16 or 17, wherein the at least one mode of operation is: 
     a mode of operation with a locked rotor condition of a motor controlled by the three-phase power inverter; or 
     a mode of operation where a rotation speed of the motor is below a predefined threshold. 
     Example 19 
     The method of one of examples 16 to 18; 
     wherein in the at least one mode of operation in each control period the control is based on two active vector delimiting a sector indicated by a feedback angle and on two different null vectors. 
     Example 20 
     The method of example 19; 
     wherein said controlling comprises employing, in the at least one mode of operation, in each control period; 
     four different sequences of the two active vectors and the two null vectors; and 
     each sequence including one of the two active vectors and one of the two null vectors. 
     Example 21 
     The method of example 20; 
     wherein each sequence includes the one of the two active vectors followed by the one of the two null vectors. 
     Example 22 
     The method of example 20 or 21; 
     wherein said controlling comprises controlling the three-phase power inverter in each control period according to a control scheme {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt; a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 0 , where {right arrow over (V)} a , {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 23 
     The method of example 19; 
     wherein said controlling comprises employing, in the at least one mode of operation, in each control period; 
     a first sequence including one of the active vectors followed by two different null vectors; and 
     a second sequence including the other one of the two active vectors followed by two different null vectors. 
     Example 24 
     The method of example 23; 
     wherein the first sequence is one of {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7  or {right arrow over (V)} a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0 ; and 
     the second sequence is one of {right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0  or {right arrow over (V)} b -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 , where {right arrow over (V)} a , {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 25 
     The method according to example 23 or 24; 
     wherein said controlling comprises employing one of the active vectors between the first sequence and the second sequence. 
     Example 26 
     The method according to example 19; 
     wherein said controlling comprises employing, in the at least one mode of operation, in each control period; 
     two different sequences of two vectors; 
     each sequence including one of the two active vectors and one of two null vectors; and 
     one sequence including one of the two active vectors followed by two different null vectors. 
     Example 27 
     A computer program comprising a program code, which, when executed on one or more processors, causes execution of the method of any one of examples 16 to 26. Causing execution means in particular that the one or more processors act as controller controlling execution of the method. 
     Example 28 
     A computer program comprising a program code for controlling a three-phase power inverter; 
     the three-phase power inverter comprising three half-bridges each comprising two switches and two diodes coupled in anti-parallel to the switches as power devices, which program code, when executed on one or more processors, causes using field-oriented control via space vector pulse width modulation; and 
     in at least one mode of operation, controlling the three-phase power inverter such that in each control period of the space vector pulse width modulation four of the power devices take turns in bearing a full current during application of a null vector; 
     null vectors being vectors where all three half-bridges are controlled in the same manner; and 
     wherein a full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     Example 29 
     A tangible storage medium storing the computer program of example 27 or 28. 
     Example 30 
     A device for controlling a three-phase power inverter; 
     the three-phase power inverter comprising three half-bridges each comprising two switches and two diodes anti-parallel to the switches as power devices; 
     the device comprising: 
     means for using field-oriented control via space vector pulse width modulation; and 
     means for controlling, in at least one mode of operation, the three-phase power inverter such that in each control period of the space vector pulse width modulation four of the power devices take turns in bearing a full current during application of a null vector; 
     null vectors being vectors where all three half-bridges are controlled in the same manner; and 
     wherein a full current is an absolute current value of a maximum phase current among three phase currents of the three-phase power inverter. 
     Example 31 
     The device of example 30; 
     wherein the at least one mode of operation is a mode of operation with a locked rotor condition of a motor controlled by the three-phase power inverter; or 
     a mode of operation where a rotation speed of the motor is below a predefined threshold. 
     Example 32 
     The device of example 30 or 31; 
     wherein in the at least one mode of operation, in each control period the control is based on two active vector delimiting a sector indicated by a feedback angle and on two different null vectors. 
     Example 33 
     The device of example 32; 
     wherein said means for controlling comprises means for employing, in the at least one mode of operation, in each control period: 
     four different sequences of the two active vectors and the two null vectors; and 
     each sequence including one of the two active vectors and one of the two null vectors. 
     Example 34 
     The device of example 33; 
     wherein said means for controlling comprises means for controlling the three-phase power inverter in each control period according to a control scheme {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} b -&gt;{right arrow over (V)} 0 , where {right arrow over (V)} a , {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 35 
     The device of example 32; 
     wherein said means for controlling comprises means for employing, in the at least one mode of operation, in each control period: 
     a first sequence including one of the active vectors followed by two different null vectors; and 
     a second sequence including the other one of the two active vectors followed by two different null vectors. 
     Example 36 
     The device of example 35; 
     wherein the first sequence is one of {right arrow over (V)} a -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7  or {right arrow over (V)} a -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0 , and the second sequence is one of {right arrow over (V)} b -&gt;{right arrow over (V)} 7 -&gt;{right arrow over (V)} 0  or {right arrow over (V)} b -&gt;{right arrow over (V)} 0 -&gt;{right arrow over (V)} 7 , where V, {right arrow over (V)} b  are the two active vectors, {right arrow over (V)} 7  is a first null vector, and {right arrow over (V)} 0  is a second null vector. 
     Example 37 
     The device according to example 35 or 36; 
     wherein said means for controlling comprises means for employing one of the active vectors between the first sequence and the second sequence. 
     Example 38 
     The method according to example 32; 
     wherein said means for controlling comprises means for employing, in the at least one mode of operation, in each control period: 
     two different sequences of two vectors; 
     each sequence including one of the two active vectors and one of two null vectors; and 
     one sequence including one of the two active vectors followed by two different null vectors. 
     Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.