Patent Publication Number: US-9425511-B1

Title: Excitation method of coaxial horn for wide bandwidth and circular polarization

Description:
BACKGROUND 
     1. Field 
     This invention relates generally to a wide bandwidth, narrow beam coaxial antenna feed horn and, more particularly, to a wide bandwidth, coaxial antenna feed horn that includes a tapered dielectric at the horn aperture for impedance matching to free space and/or a multi-layered dielectric member that allows propagation of a TE 11  sum mode and a TE 12  difference mode starting at the same cut-off frequency, where polarization may be linear or circular. 
     2. Discussion 
     For certain communications applications, it is desirable to have a broadband system, namely, operation over a relatively wide frequency range, typically greater than 1.5:1. In some reflector based systems, it is desirable to have a feed with a small foot print, making it suitable for illuminating very low focal length to diameter ratios reflector lens. 
     In certain communications systems, signal tracking between the receiver and transmitter is achieved with the use of a sum and difference pattern. A sum pattern presents a broadside peak radiation pattern, while a difference pattern provides a broadside null radiation pattern. In this case, two electromagnetic propagation modes, the transverse-electric (TE) modes (TE 11 , TE 12 ) in the feed horn are needed to realize a sum and difference within the same frequency range. In general, the TM 00  mode is used for linear polarization. System performance requirements may call for a large instantaneous RF bandwidth and a small physical footprint, to name a few. 
     A critical element to achieve the signal tracking feature, while meeting system specifications is the feed antenna. To meet size constraints, a smaller aperture size is usually desired, such as that of a coaxial horn antenna. However, its cut-off frequency of the TE 12  difference mode is twice the cut-off frequency of the TE 11  sum mode, where the cut-off frequency of a particular mode is the lowest frequency that the mode can propagate. It is known in the art to load such a feed horn with a dielectric to lower the cut-off frequency of a particular mode. In addition to realizing the necessary modes for generating the sum and difference mode, ample signal from the feed horn must be transmitted/received. Namely, for a small aperture relative to the operating wavelength feed horn, there exists a significant impedance mismatch between the dielectric and free space resulting in significant signal loss. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is an isometric view of a coaxial antenna feed horn; 
         FIG. 2  is a cross-sectional view of the feed horn shown in  FIG. 1 ; 
         FIG. 3  is a cut-away, bottom isometric view of the feed horn shown in  FIG. 1 ; 
         FIG. 4  is a cross-sectional view of a coaxial antenna feed horn including multiple dielectric layers; 
         FIG. 5  is an illustration showing circularly polarized excitation for a TE 11  sum mode; 
         FIG. 6  is an illustration showing circularly polarized excitation for a TE 12  difference mode; and 
         FIG. 7  is a representative directivity plot with elevation angles (degrees) represented on the horizontal axis and directivity (dB) on the vertical axis showing a TE 11  sum mode circular polarization pattern and a TE 12  difference mode circular polarization pattern. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     The following discussion of the embodiments of the invention directed to a broadband coaxial antenna feed horn is merely exemplary in nature, and is in no way intended to limit the invention or its applications or uses. 
       FIG. 1  is an isometric view,  FIG. 2  is a cross-sectional view and  FIG. 3  is a cut-away, bottom isometric view of a coaxial antenna feed horn  10  having appropriate dimensions for a particular wide bandwidth frequency band, for example, 21-51 GHz. The horn  10  includes a dielectric substrate  12 , such as Rogers Duroid, having, for example, a relative dielectric constant ∈ r =3. A conductive finite ground plane  14 , such as copper, is deposited on a top surface of the substrate  12  and is in electrical contact with an outer cylindrical ground conductor  16 , such as copper, defining a cylindrical chamber  36  therein. The conductor  16  includes a tapered portion  18  defining an aperture  22  of the horn  10  opposite to the substrate  12 , as shown. A circular ground plane  20  is in electrical contact with the outer conductor  16  proximate the aperture  22 , as shown. The ground plane  20  can be any applicable size and/or shape for a particular embodiment, and can be electrically coupled to the conductor  16  at any location along its length. Further, it is noted that the ground plane  20  can be eliminated in some embodiments. 
     An embedded conductor  24  is provided within the chamber  36  and is coaxial with the ground conductor  16 , where the embedded conductor  24  includes a lower conical section  26 , a middle cylindrical section  28  and a tapered section  30  extending through the aperture  22 . A dielectric member  32  is provided within the chamber  36  between the embedded conductor  24  and the outer conductor  16  and includes a tapered end section  34  surrounding the tapered section  30  and extending from the aperture  22 . A series of four microstrip feed lines  38  positioned at 90° relative to each other are deposited on a bottom surface of the substrate  12  opposite to the ground plane  14 . In this non-limiting embodiment, four independent microstrip lines  40  attached to the feed lines  38  and extends through the substrate  12  to be electrically attached to a cylindrical feed line transition member  42  that is electrically attached to a lower end of the conical section  26  of the embedded conductor  24 . The conical section  26  provides a microstrip-to-coaxial mode transformer that allows a signal on the microstrip feed lines  38  propagating in the microstrip mode to be converted to a coaxial transmission mode. The conductive material discussed herein can be any suitable conductor, such as copper, where the embedded conductor  24  can be a solid piece or be hollow. 
     The tapered section  34  of the dielectric member  32  provides a transition for impedance matching between the aperture  22  of the feed horn  10  and free space. It is typically desirable to provide a transition of the tapered section  34 , which makes it longer, to provide the best impedance matching to free space. In one non-limiting embodiment for the frequency band mentioned above, the dielectric member  32  can be Teflon having a dielectric constant of ∈ r =2.1, and the tapered section  34  has a length of about 0.63 in. The conical section  26  provides impedance matching between the microstrip lines  38  and  40  and the embedded conductors  28 ,  36 . Further, excitation signals applied to the microstrip lines  38  are phased to excite the TE 11  sum mode in the horn  10 , which generates a circularly polarized sum pattern. 
     The dielectric member  32  extends the length of the horn  10  and is homogeneous, i.e., has the same dielectric constant from top to bottom. In this design, the TE 12  difference mode cut-off frequency is still above the TE 11  sum mode cut-off frequency. In order to reduce the cut-off frequency of the TE 12  difference mode to be the same as that of the TE 11  sum mode so that they propagate within the desired frequency range for signal tracking, the present invention proposes providing a TE 12  difference mode excitation signal to the antenna feed horn  10  and provide a transition in the dielectric constant of the dielectric  32  to reduce the cut-off frequency of the TE 12  difference mode. By loading the feed horn with a relatively higher dielectric material, the cut-off frequency for the TE 12  difference mode can be lowered to the cut-off frequency of the TE 11  sum mode, thus allowing both modes to propagate at the same time and at the same frequency, although in axially different locations. 
       FIG. 4  is a cross-sectional view of a coaxial antenna feed horn  50  showing this embodiment that is similar to the feed horn  10 , where like elements are identified by the same reference number. In this design, the dielectric member  32  is replaced with a plurality of dielectric layers with different dielectric constants ∈ r  from the bottom of the feed horn  50  to the top of the feed horn  50  to provide impedance matching. For example, a dielectric layer  52  is provided at the bottom of the feed horn  50  within the conductor  16  and has a dielectric constant ∈ r  that allows propagation of the TE 11  sum mode, such as Teflon having a constant ∈ r =2.1, where the TE 11  sum mode is generated by the excitation signal applied to the microstrip lines  38 . A plurality of other dielectric layers are provided on top of the dielectric layer  52  in ascending order of dielectric constant ∈ r  to provide impedance matching between the layers in this non-limiting embodiment. In this particular design, a dielectric layer  54  is provided on top of the dielectric layer  52  and has a larger dielectric constant ∈ r  than the dielectric layer  52 , a dielectric layer  56  is provided on top of the dielectric layer  54  and has a larger dielectric constant ∈ r  than the dielectric layer  54 , and a dielectric layer  58  is provided on top of the dielectric layer  56  and includes a tapered section  60  extending out of the aperture  18 , where the dielectric layer  58  has a larger dielectric constant ∈ r  than the dielectric layer  56 . The dielectric layer  58  also has the proper dielectric constant ∈ r  that allows propagation of the TE 12  difference mode, such as ∈ r =6. It is noted, that the TE 11  sum mode propagates in and above the lines  40 , and the orthogonal TE 12  difference mode propagates in and above the layer  58 . 
     In one non-limiting embodiment shown merely for illustrative purposes, the dielectric layer  54  is fused silica having a dielectric constant ∈ r =3, the dielectric layer  56  is boron nitride having a dielectric constant ∈ r =4 and the dielectric layer  58  is beryllium oxide having a dielectric constant ∈ r =6. Further, also by way of a non-limiting example, the dielectric layer  52  can be 0.13″, the dielectric layer  54  can be 0.248″, the dielectric layer  56  can be 0.193″ and the cylindrical portion of the dielectric layer  58  below the aperture  18  can be 0.176″. 
     For this embodiment, an excitation signal needs to be applied to the horn  50  to generate the TE 12  difference mode and needs to be applied in the area of the dielectric layer  58 , which has the dielectric constant ∈ r  that allows the TE 12  difference mode to propagate in the horn  50  at the lower cut-off frequency. This signal can be applied in any suitable manner to the horn  50 . As a general representation of this, an electrical probe  44  is shown proximate the dielectric layer  58  to which the TE 12  difference mode excitation signal is provided. 
     In order to obtain the TE 11  sum propagation mode, a uniform amplitude phase changing excitation signal is applied to the microstrip lines  38 . For example,  FIG. 5  is an illustration  64  showing electrical terminals  66  at positions 0°, 90°, 180° and 270° around an outer conductor  68  representing the lines  40  to which the TE 11  sum propagation mode excitation signal is selectively applied in rotation. 
     In order to obtain the TE 11  sum propagation mode and the TE 12  difference propagation mode, a uniform amplitude phase changing excitation signal is applied to the microstrip lines  38  and  44 . For example,  FIG. 6  is an illustration  70  showing electrical terminals  72  at positions 0°, 90°, 180° and 270° around an outer conductor  74  representing the microstrip lines  40 . In order to obtain the TE 12  difference propagation mode, a constant amplitude phase changing excitation signal is provided to  70  at each of the electrical terminals  72 . The relative phase difference at each electrical terminal  72 , in a counter clockwise fashion are 0°, 90°, 180°, 270°, 0°, 90°, 180°, 270. 
       FIG. 7  is a representative directivity plot with elevation angles (degrees) represented on the horizontal axis and directivity (dB) on the vertical axis showing a TE 11  sum mode circular polarization. Particularly, plot line  84  is the TE 11  sum antenna pattern and plot line  86  is the TE 12  difference antenna pattern. 
     The foregoing discussion disclosed and describes merely exemplary embodiments of the present invention. One skilled in the art will readily recognize from such discussion and from the accompanying drawings and claims that various changes, modifications and variations can be made therein without departing from the spirit and scope of the invention as defined in the following claims.