Patent Publication Number: US-7593482-B2

Title: Wireless communication system with hardware-based frequency burst detection

Description:
CROSS REFERENCE TO RELATED PATENT APPLICATIONS 
     This patent application is related to the U.S. Patent Application entitled “Highly Integrated Radio-Frequency Apparatus and Associated Methods”, inventors Navdeep S. Sooch and G. Tyson Tuttle, Ser. No. 10/426,042 filed Apr. 29, 2003, the disclosure of which is incorporated herein by reference in its entirety. 
     This patent application is related to the U.S. Patent Application entitled “Wireless Communication System and Method With Frequency Burst Acquisition Feature Using Autocorrelation”, inventors Vis et al., (Ser. No. 10/955,584 and filed the same day as this application) the disclosure of which is incorporated herein by reference in its entirety. 
     This patent application is related to the U.S. Patent Application entitled “Wireless Communication System and Method With Frequency Burst Acquisition Feature Using Autocorrelation”, inventors Liang et al., (Ser. No. 10/954,791 and filed the same day as this application) the disclosure of which is incorporated herein by reference in its entirety. 
     TECHNICAL FIELD OF THE INVENTION 
     The disclosures herein relate generally to highly integrated wireless communication systems and more particularly to the reduction of digitally generated noise in such systems. 
     BACKGROUND 
     Mobile wireless communications systems, such as GSM/GPRS (Global System Mobile/General Radio Packet Service) systems, include base stations that can communicate with large numbers of mobile stations. The base stations periodically transmit a reference signal known as a frequency burst (FB) on a broadcast control channel (BCCH) to enable the mobile stations to synchronize with the base stations. The purposes of the frequency burst are twofold, namely frequency calibration and coarse timing. First, with respect to frequency calibration of the mobile station, the frequency burst enables correction of the frequency offset or error between the local carrier frequency of the mobile station and the carrier frequency of the base station. Second, with respect to coarse timing, approximate time alignment with the beginning of the frequency burst is performed so that a fine tuning can be conducted in the later acquisition of a sync burst (SB) from the base station. 
     These mobile communication systems typically include radio frequency front ends with analog circuitry that is very noise and interference sensitive. In some applications, the RF circuitry may need to detect signals as small as a few micro-volts in amplitude. If this RF circuitry encounters noise, it is possible that receive performance will be substantially degraded. These systems often employ signal processing circuitry such as a digital signal processor (DSP) to search for the frequency burst (FB) signal and a microcontroller unit (MCU) to govern the overall operation of the system. The digital circuitry of the DSP and MCU produces digital signals with relatively small rise and fall times, or with fast transitions or sharp edges. The large size and the high gate count of the DSP and MCU are factors which contribute to the substantial amount of noise that may be produced in the vicinity of the DSP and MCU. Unfortunately, it is possible that this digitally generated noise may significantly interfere with the operation of radio frequency circuitry of the system. 
     What is needed is an apparatus and methodology which achieves frequency burst acquisition without subjecting radio frequency circuitry to substantial amounts of noise. 
     SUMMARY 
     Accordingly, in one embodiment, a method is disclosed for operating a mobile telephone apparatus including radio frequency (RF) circuitry. The method includes activating the RF circuitry to receive RF signals. The method also includes processing, by first digital circuitry and second digital circuitry, information derived from the received RF signals. The first digital circuitry is configured to locate a frequency burst in the received RF signals. The second digital circuitry is configured to perform processes other than frequency burst location. The method further includes inactivating the second digital circuitry at times when the RF circuitry is active to receive RF signals. 
     In another embodiment, a mobile telephone apparatus is disclosed that includes RF circuitry that receives RF signals when active. The apparatus also includes first digital circuitry that is coupled to the RF circuitry. The first digital circuitry is configured to locate a frequency burst in received RF signals. The apparatus also includes second digital circuitry that is coupled to the first digital circuitry. The second digital circuitry is configured to perform an activity other than frequency burst location and is inactive when the RF circuitry is active. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The appended drawings illustrate only exemplary embodiments of the invention and therefore do not limit its scope, because the inventive concepts lend themselves to other equally effective embodiments. 
         FIG. 1  is a block diagram of one embodiment of the disclosed wireless communication system, for example a mobile telephone apparatus. 
         FIG. 2  is a graph of a typical autocorrelation (ACOR) ratio profile when an FB is detected. 
         FIG. 3  is a state diagram of an FB location state machine of the FB search hardware with CW detection capability. 
         FIG. 4  is a state diagram of an FB location state machine of the FB search hardware without CW detection capability. 
         FIG. 5  is a flowchart illustrating the operation of a filter bank of the frequency offset estimation process employed by the disclosed system. 
         FIG. 6  is a flowchart illustrating the operation of a single filter of the frequency offset estimation process employed by the disclosed system. 
         FIG. 7A-7B  together form a flowchart summarizing the operation of the disclosed wireless communication system. 
     
    
    
     DETAILED DESCRIPTION 
     When RF circuitry, such as receiver circuitry, is integrated with signal processing circuitry on a common integrated circuit, a problem may be encountered with noise from digital circuitry within the signal processing circuitry interfering with the RF circuitry. This can be caused by the very close proximity of the RF circuitry with respect to the signal processing circuitry. To enable reliable acquisition of the frequency burst (FB) in the received signal, this noise problem must be addressed. 
     One approach to this noise problem is to inactivate the noise producing digital circuitry, such as the DSP and MCU described below, during those times when the RF circuitry is activated. Unfortunately, this means that the digital circuitry can not determine the FB from the received signals in real time. One option is to use a large buffer to store the received radio data during the receive time and then the data can be processed later when the radio circuitry is inactivated and the digital circuitry is activated. However, this approach has several drawbacks. For example, the size of the buffer needed may be very large and precious DSP cycles may be consumed during the narrow window of time when the DSP is allowed to operate, i.e. when the RF circuitry is inactivated. The approach now described solves these problems and achieves real time processing of received signals to determine the location of the FB in one embodiment. 
     One embodiment of the disclosed wireless communication system includes RF circuitry to receive and transmit radio frequency signals. The system also includes digital circuitry that processes various signals in the system as described below. The communication system is configured such that the RF circuitry can be activated and inactivated, i.e. so that the RF circuitry can exhibit an active state during which it is receiving RF signals and an inactive state during which it is in standby, as described in more detail below. The noise producing digital circuitry, for example the DSP and MCU described below, is also configured such that it can be activated and inactivated, i.e. so that the digital circuitry can exhibit an active state or an inactive state. It is noted that inactivating a particular circuit may be accomplished by powering the circuit down, disabling the circuit, placing the circuit in standby or otherwise inhibiting the operation of the circuit. 
     In one embodiment, the noise producing digital circuitry is caused to be inactive when the RF circuitry is active to receive the RF signals that are needed for determining the location of the FB. The system includes a dedicated frequency burst (FB) search hardware circuit that is active during reception of RF signal samples at the same time that the noise producing digital circuitry is inactive. The FB search hardware circuit is digital circuitry; however, it exhibits relatively low noise in comparison with other noise producing digital circuitry of the system such as the DSP and MCU. In one particular embodiment, the RF circuitry is active and the dedicated FB search hardware circuit is active to receive and process RF signal samples while other noise producing digital circuitry such as the DSP and MCU are held inactive. Once the dedicated FB search hardware circuit finds the FB or the below-described FB search window closes, the noise producing digital circuitry such as the DSP and MCU can be reactivated. In such embodiment, this digital circuitry is inactivated or turned off to reduce the impact of digital noise on the RF circuitry when the RF circuitry is active. This approach is referred to as time domain isolation in which noise producing digital activities are isolated in time from noise sensitive RF reception activities. The RF circuitry may be inactivated before resuming or reactivating the DSP and MCU digital circuitry, but that is not required in all applications. The time during which the RF circuitry is active is controlled by a timer so that the time during which the RF circuitry is active is segmented or isolated in time from the majority of the digital switching. Once a time domain isolation (TDI) window, described below, has passed, the RF circuitry has no further need to avoid noise. Thus, the RF circuitry may be left in an active state and its output may be ignored after passage of the TDI window. It is also possible that the RF circuitry remain powered on at all times and that the RF circuitry is utilized at times when noise from digital circuitry is not present, i.e. when the digital circuitry is inactive. In an alternative embodiment, signals from the RF circuitry can be ignored at times during which the digital circuitry, such as the DSP and MCU, is active depending on the particular application. 
       FIG. 1  is a block diagram of one embodiment of the disclosed wireless communication system  100 , for example a mobile telephone apparatus. A high level discussion of the operation of system  100  is presented first followed by a more detailed discussion of the subsystems forming system  100 . System  100  includes RF circuitry  115 , namely a transceiver subsystem, and signal processing circuitry  190 , namely a baseband section or subsystem as shown. In one embodiment, RF circuitry  115  and signal processing circuitry  190  are situated on a common semiconductor chip  102 , or otherwise co-packaged such as in a multi-chip module (MCM) or in the same EMI cavity. In more detail, the RF circuitry  115  of system  100  includes an antenna  105  which is coupled to a surface acoustic wave (SAW) filter  110 . In this embodiment SAW filter  110  is configured to select a radio frequency (RF) signal from one of four bands, namely the GSM 850, E-GSM 900, DCS 1800, and PCS 1900 MHz bands. Those skilled in the art will appreciate that the system is readily adapted to operate on bands other than those given in this particular example. In actual practice, antenna  105  and filter  110  may be located external to semiconductor chip  102 . 
     SAW filter  110  is coupled to receiver  120  as illustrated in  FIG. 1 . For simplicity, the transmitter portion of transceiver  115  is omitted in  FIG. 1 . Signal processing circuitry  190  includes a micro-controller unit (MCU)  125  that controls the operation of subsystems of wireless communication system  100 . MCU  125  is illustrated as having an automatic frequency control (AFC) output  225 A which is coupled to an input of a digitally controlled crystal oscillator (DCXO)  130  to control the operating frequency of the DCXO. However, the line connecting AFC output  225 A to DCXO  130  represents a logical connection. In actual practice the MCU  125  writes frequency adjustment information to an MCU peripheral (not shown) and that MCU peripheral communicates the information to an RF control port (not shown) of RF circuitry  115 . The AFC output may be located in either MCU  125  or DSP  200 . A crystal  135  is coupled to DCXO  130  to provide a time base reference to the DCXO. DCXO  130  provides a reference frequency signal to a frequency synthesizer  140  which is coupled thereto. The output of frequency synthesizer  140  is coupled via a 0°/90° phase shifter  145  to an analog mixer  150 . Phase shifter  145  shifts the phase of the signals provided thereto by either 0 degrees or 90 degrees. Thus, when the output signal of shifter  145  is combined with the received signal in analog mixer  150 , the signal is separated into in-phase (I) and quadrature (O) components. 
     The in phase (I) and quadrature (O) components are converted from analog signals to digital signals by analog to digital converters (ADCs)  155  and  160 , respectively. The resultant digital I and digital Q signals are then provided to a digital mixer  165  which is coupled to a low intermediate frequency block  170  to down-convert the digital I and Q signals to baseband frequencies. A dual mode filter  175  is coupled to digital mixer  165  to separate the I and Q signals at baseband from adjacent signals outside of the pass bands of the filter. In conventional designs, a filter is normally centered on the nominal carrier frequency where the mobile station is operating. To include the desired signal within the pass band, the bandwidth of the filter needs to be large enough to cover the worst cases with maximum positive and maximum negative frequency offset. By doing so, an inherent disadvantage is that it reduces isolation from adjacent channel signals, which can leak into the pass band and act as interference sources. The GSM/GPRS specification defines an FB signal to be positioned at approximately 67.7 kHz beyond the nominal carrier frequency. In the disclosed embodiment, an FB search custom filter  175 A is employed when frequency offset can be relatively large. The center frequency of this custom filter  175 A is chosen to be between the nominal carrier frequency and the corresponding FB frequency, and the bandwidth of the filter is narrower than for the reception of other types of GSM/GPRS burst, such as synchronization burst (SB) and normal burst (NB). Hence, the custom filter  175 A is less vulnerable to adjacent interference and also passes the desired FB signal, even if the frequency error is relatively large. In the embodiment described above, the custom filter  175 A is an asymmetric complex filter, which is more expensive to implement in comparison with a symmetric real filter. A preferable alternative filter  175 A is a symmetric, real-valued narrow-band custom filter wherein its pass-band is shifted by biasing the synthesizer frequency away from the nominal value. In this way, the resulting filter is centered on the biased synthesizer frequency and thus the asymmetric filter described above can be implemented as a symmetric real filter plus a synthesizer frequency bias. 
     More specifically, during initial acquisition when frequency offset is relatively large, an FB search custom filter  175 A is switched into use, or otherwise activated. Additionally, the local frequency generated from DCXO  130  is intentionally biased away from the nominal value on which the mobile station is operating. The bias value and the bias direction are pre-defined and directly related with the design of the FB custom filter  175 A. Once frequency synchronization is achieved, channelization filter  175 B is switched into use during subsequent FB searches and the frequency bias is not applied. During reception of synchronization burst (SB) and normal burst (NB), the channelization filter  175 B is switched into use, or otherwise activated, as filter  175 . In one embodiment, when FB search filter  175 A is active, channelization filter  175 B is inactive, and vice versa. One FB search filter  175 A that may be employed as filter  175  is a combination of two cascaded filters working at different sampling rates. FB search filter  175 A has a bandwidth that is narrower than the bandwidth of channelization filter  175 B. Channelization filter  175 B is a low pass filter exhibiting a bandwidth wider than narrow band FB filter  175 A, namely a bandwidth appropriate for normal reception of GSM voice and data signals. 
     The I and Q outputs of filter  175  are coupled to RF receiver interface  180 . It is noted that the I and Q outputs of filter  175  are also outputs of transceiver subsystem  115 . To minimize noise in one embodiment, only the storage portion of RF receiver (RF RX) interface  180  is active when RF circuitry  115  is in use. The storage portion includes writing logic (not shown) and random access memories (RAMs) of an FB circular buffer  182 . Circular buffer  182  is configured to store I and Q samples that are supplied thereto by receiver  120  of RF circuitry  115 . However, when DSP  200  is active, the entire RF RX interface  180  is active. 
     I and Q outputs  180 A- 180 B of RF receiver interface  180  are coupled to frequency burst (FB) search hardware  185  (FBSHW) which determines the location of the frequency burst (FB) when receiver  120  is first set to a particular frequency channel. I and Q outputs  180 A- 180 B are associated with the above described storage portion of RF RX interface  180  that is active when RF circuitry  115  is in use. FB search hardware  185  is located in baseband subsystem  190  which is also designated as signal processing circuitry  190 . Other I and Q outputs  180 C- 180 D of RF receiver interface  180  are coupled to a digital signal processor (DSP)  200  which, when active, executes a frequency offset estimation process  205  to adjust the frequency of receiver  120  when a search for the FB is conducted. These I and Q outputs  180 C- 180 D are active when DSP  200  is active. In one embodiment, FB search hardware  185  is dedicated to searching for FB in real time when RF circuitry  115  is active while DSP  200  and MCU  125  are held inactive to control noise as will be described in more detail below. In one embodiment, RF RX interface  180  and FB search hardware  185  are both digital circuitry which exhibit relatively low noise as compared with the relatively high noise exhibited by other digital circuits such as DSP  100  and MCU  125  when they are active. The relatively low noise level generated by FBSHW  185  is sufficiently low that it does not significantly interfere with the operation of receiver  120 . 
     FB search hardware  185  includes an auto-correlation (ACOR) computation unit  195  to which the I and Q outputs of receiver interface  180  are coupled. Any DC offset which the I and Q signals may contain can be removed in ACOR unit  195  based on the value in DC offset register  246  as programmed by MCU  125 . One purpose of FB search hardware  185  is to distinguish between a desired FB signal on a particular channel and an interferer. On the GSM/GPRS broadcast control channel (BCCH), each FB lasts for one time slot while other types of bursts generated from quasi-random data are transmitted prior to and after the FB. Thus, a relatively sharp or relatively high peak appears in the magnitude of the ACOR coefficients of the received signal when it reaches the end of an FB. Once the position of that ACOR peak is determined, coarse timing of the FB is achieved. An interferer may be a so-called “continuous wave” (CW) signal or other similar undesired narrowband signal associated with existing analog cellular systems or other digital systems. It is noted that for a CW signal or other similar undesired narrowband signal, such a signal lasts much longer than an FB. By observing the profile of ACOR values of the received signal, the disclosed FB search hardware  185  can distinguish between the desired GSM signal and a narrowband interferer. 
     FB search hardware  185  includes an FB location state machine section  210  that performs profile matching to find the sharp peak in the ACOR function values, i.e. the ACOR coefficients, mentioned above. FB location state machine section  210  includes an FB location state machine with CW detection  215  and an FB location state machine without CW detection  220  to determine the location of the FB from information provided thereto by ACOR computation unit  195 . Once the FB is located, an automatic frequency correction (AFC) unit  225  of a synchronization state machine  230  in MCU  125  uses this result to adjust the DCXO programming value. This indirectly changes the carrier frequency of RF receiver  115  by making adjustments to a 26 MHz clock associated with frequency synthesizer  140 . The disclosed methodology can compensate for relatively large carrier frequency offsets. In practice, this means that less expensive crystals  135  may be employed as frequency references in system  100 . Such crystals tend to have wider tolerance and thus tend to exhibit larger offsets for which the disclosed method and apparatus can compensate. 
     Synchronization state machine  230  of MCU  125  includes an FB scheduling unit  235  which controls the timing and duration of the search window during which receiver  120  is activated to listen for FB on the desired channel. As will be discussed in more detail below, it is noted that DSP  200  and MCU  125  are not always activated. Rather, to reduce digital noise that could impact reception, when receiver  120  is active, DSP  200  and MCU  125  are inactive. Then later, after receiver  120  is deactivated, DSP  200  and MCU  125  can be activated. This is a type of time domain isolation (TDI) wherein RF activities are generally isolated in time from digital processing activities. Time domain isolation is described in the copending patent U.S. patent application entitled “Highly Integrated Radio-Frequency Apparatus and Associated Methods”, inventors Navdeep S. Sooch and G. Tyson Tuttle, Ser. No. 10/426,042 filed Apr. 29, 2003, the disclosure of which is incorporated herein by reference in its entirety. A time domain isolation window (TDI window) is a period of time during which RF activity is isolated from digital activity that is performed at other times. A system timer unit (STU)  240  is coupled to MCU  125  to receive system timing instructions therefrom. In one embodiment STU  240  is always on. STU  240  includes a power down (POWER DOWN) output  240 A which is coupled to transceiver  115  so that receiver  120  can be activated or inactivated. In one embodiment, STU  240  generates a POWER DOWN signal which controls when the RF circuitry is active and inactive. It is noted that the RF circuitry  115  can include not only receiver circuitry, but also transmitter circuitry. The POWER DOWN signal can be used to control the activation and inactivation of both receiver and transmitter circuitry. STU  240  also includes an FB_Search_Enable output  240 B which is coupled to FB search hardware  185  so that hardware  185  can be instructed that an FB search window is open and that an FB search is to be commenced. In one embodiment, receiver  120  is active during the search window. In another embodiment, receiver activation is independent of the FB search window. This means that the FB search window may open before or after activation of the receiver, and may close before or after the receiver is deactivated. 
     More detail is now provided with respect to the subsystems and other components of wireless communication system  100 . As mentioned above, FB search hardware  185  includes an autocorrelation (ACOR) computation unit  195  that determines the autocorrelation results or values with respect to the signal received by receiver  120  in real time when receiver  120  is active and DSP  200  and MCU  125  are inactive. A sharp peak in the magnitude of the ACOR results or values of the received signal indicates that an FB has been received. More specifically, this sharp peak corresponds to the end of an FB. Once the position of this peak is determined, coarse timing of the FB is accomplished. The ACOR results, i.e. the ACOR Ratios or values, can be determined by using either Equation 1 or Equation 2 below. In one embodiment, ACOR computation unit  195  calculates the magnitude of the autocorrelation value by using 144 baud-rate symbols in a sliding window. A delay lag of 8 symbols is used and the sliding window advances 9 symbols at a time. More particularly, nine symbols are averaged and every nine symbols the oldest of the nine symbol averages in a collection of sixteen is replaced by the newest average, and the ACOR of the resulting 144 bits is performed as indicated in Equation 2 below. In one embodiment, a post-normalized ACOR value, Ratio post [i], is determined according to Equation 1: 
                       Ratio   post     ⁡     [   i   ]       =                ∑     k   =     i   ×   9         k   =       i   ×   9     +   143         ⁢       y   ⁡     [   k   ]       ·       y   *     ⁡     [     k   -   8     ]                2                ∑     k   =       i   ×   9     -   4         k   =       i   ×   9     +   139         ⁢            y   ⁡     [   k   ]            2            2               Equation   ⁢           ⁢   1               
wherein 8 is the autocorrelation delay of 8 GSM symbols;
 
wherein Ratio post [i] is computed at 1/9 of the GSM symbol rate, the symbol index “k” being advanced by 9 each time. The summation for the numerator is performed in a time window k=[i*9, i*9+143] which contains 144 symbols. The summation for the denominator is performed in a time window k=[i*9−4, i*9+139] which also contains 144 symbols. The term, y[k], is the baud-rate sampled received signal.
 
     In one embodiment, to improve the statistics of ACOR value in a fading environment, a pre-normalized ACOR value, Ratio[i], is determined according to Equation 2: 
                     Ratio   ⁡     [   i   ]       =              1   144     ⁢       ∑     k   =     i   ×   9         k   =       i   ×   9     +   143         ⁢         y   ⁡     [   k   ]       ·       y   *     ⁡     [     k   -   8     ]                  y   ⁡     [   k   ]       ·     y   ⁡     [     k   -   8     ]                         2             Equation   ⁢           ⁢   2               
The term, y[k], is again the baud-rate sampled received signal. In this particular embodiment, the FB search conducted by the ACOR computation unit  195  determines the autocorrelation value at a fixed time delay of 8 symbols according to this equation. Time delays less than or greater than 8 symbols can also produce acceptable results provided the time delay is not so large or small that performance is substantially degraded. When the time delay is set to 8 symbols to perform the calculation indicated by Equation 2 for the pre-normalized ACOR value, Ratio[i], 9 symbols are averaged. For every 9 symbols, the oldest of the 9 symbol averages in a collection of 16 is replaced by the newest average and the ACOR value for the resulting 144 bits is determined. Thus the computation is performed every 9 samples. The initial average is over 9 samples and thus there are 8 clocks to perform the average. In one embodiment, ACOR computation unit  195  may employ a look-up table (not shown) for the pre-normalization computation, rather than applying a divide operation.
 
     The values, Ratio[i], are supplied to the frequency burst FB location state machine section  210  so that state machine section  210  can determine if an FB has been found. Location state machine section  210  includes an FB location state machine with CW detection  215  that determines if a signal is an FB or a CW signal. Location state machine section  210  also includes an FB location state machine  220  without CW detection. FB search hardware  185  is an internally programmable device which acts as a peripheral of MCU  125 . MCU  125  programs the FB search hardware with DC offset register  246  for its ACOR computation unit  195 , and programs it with mode control register  244  to select between state machine  215  and state machine  220 , and programs it with parameter register  242  for the selected state machine. It is thus seen that the MCU assists the ACOR computation unit in locating FB by these actions. Ultimately MCU  125  provides corrective information to DCXO  130  thus indirectly enabling the operating frequency of receiver  120  to be corrected. 
       FIG. 2  is a graphical representation of a typical ACOR ratio profile when an FB is detected. It is noted that this depiction is not necessarily drawn to scale and that ACOR profiles having different shapes may produce acceptable results as well. The ACOR value, Ratio[i], is shown on the y axis and time T (baud rate symbol units) is shown on the x axis. The following terms are defined with reference to  FIG. 2 . If an FB is detected at time T, then fbEndTime (the FB end time) is provided by STU  240 , an fbEndPointer (an FB end pointer) is provided by RF RX interface  180  and an fb_cnt (FB count) is provided by FB search hardware  185 . fb_cnt*9 refers to the number of baud-rate symbols from Ratio peak to the end of FB detection. MCU  125  determines an fbStartTime (an FB start time) to approximately synchronize with the time base of the transmitting base station. The fbStartPointer and fbEndPointer are pointers for FB circular buffer  182 . DSP  200  determines an fbStartPointer for its frequency offset estimation task.  FIG. 2  clearly shows a substantial peak in the autocorrelation function which is the profile associated with the occurrence of an FB. 
       FIG. 3  is a state diagram of the FB location state machine  215  with CW detection, namely narrowband interferer detection. State machine  215  is capable of distinguishing a narrowband interferer from an FB. Before discussing the operation of FB location state machine  215 , parameters employed by state machine  215  are first presented. Thresh, CW_wait, CW_fall and Minfall are parameters programmed by the MCU  125  before FB search is commenced. Ratio is the autocorrelation ratio information provided by ACOR computation unit  195 . A sequence of Ratios or values is provided to FB location state machine  210  as samples that are received and processed. CW_wait is the number of wait counts needed to claim CW detection in a FB WAIT_FOR_RISE state ( 011 ). It is noted that the names selected for the various states in the state diagrams of  FIGS. 3 and 4  are representative. These names should not be taking as being limiting and other names may be used as well to label these states. Wait counts are measured in number of Ratio [i], which is computed every 9 symbols. CW_fall is the number of wait counts needed after a Ratio peak occurs to be able to claim CW detection in a FB_WAIT_FOR_FALL state ( 101 ). With respect to the parameter Minfall, it is noted that in the WAIT_FOR_FALL state ( 101 ), after a Ratio peak has occurred, within CW_fall count, Ratio must fall for more than the “MinFall” amount in order to claim that a GSM FB has been detected. Otherwise, as seen in  FIG. 3 , the CW signal is reported as being detected. Parameters Thresh, CW_wait, CW_fall and Minfall are programmable by MCU  125  and are provided to FB location state machine  215  via parameter register  242 . Thresh is the value which the Ratio must equal or exceed to indicate that the FB has been located. CW_wait is the number of wait counts needed to claim CW detection in a FB_WAIT_FOR_RISE state. Rise_cnt is an internal counter which counts the number of Ratio[i] occurred while the state machine is in the FB_WAIT_FOR_RISE state. fb_cnt counts the number of Ratio[i] from the time when the ratio peak occurs to the time when an FB is detected. In one embodiment, FB location state machine without CW detection  220  is used, and then if an interference situation is encountered, FB location state machine  215  may used to locate the FB. 
     As seen in the state diagram of  FIG. 3 , FB location state machine with CW detection  215  commences searching for FB at state  000 . If the ACOR ratio (i.e. Ratio) is less than the threshold, Thresh, then process flow continues to the state  001  at which the state machine waits for the Ratio to exceed Thresh. State  001  is the FB_WAIT_FOR_THRESH state, a term selected for convenience. When Ratio finally exceeds or is equal to Thresh, then flow continues to state  101 , namely the FB_WAIT_FOR_FALL state ( 101 ). Ratio_fall is the difference between the ACOR ratio peak and the current ACOR ratio. Minfall is defined in the following manner. In the FB_WAIT_FOR_FALL state ( 101 ), after the ratio peak occurs, within a CW_fall count, Ratio must decrease more than the Minfall amount in order to claim that an FB is detected. CW_fall is the number of wait counts needed after a ratio peak occurs to be able to claim CW detection in a FB_WAIT_FOR_FALL state ( 101 ). After the Ratio peak has occurred, and then when Ratio_fall&gt;Minfall and fb_cnt&lt;=CW_fall, then flow continues to state  100 , namely the FB_DETECTED state. When state  100  is reached, then FB has been detected. 
     Returning to state  101  of  FIG. 3 , if Ratio_fall&lt;=MinFall and fb_cnt&gt;=CW_fall, then flow continues to state  111 , namely the FB_CW_DETECTED state. When state  111  is reached, then CW has been detected. In other words an interferer and not the desired FB has been detected. Returning to state  000 , FB_SEARCHING, if Ratio&gt;=Thresh, then flow continues to state  011 , namely the FB_WAIT_FOR_RISE state. If Ratio is then found to be less than Thresh, then flow continues to state  001  and process flow continues as discussed above. However, if at state  011 , Ratio&gt;=Thresh and Rise_cnt&gt;=CW_wait then flow continues to state  111  and a CW signal has been detected. 
       FIG. 4  is a state diagram of the FB location state machine  220  without CW detection. While other applications are possible, one application of state machine  220  is for situations when the CW carrier, i.e. narrow band interference, of another system is unlikely to be received and thus there generally is no need to distinguish between GSM&#39;s frequency burst (FB) and another system&#39;s carrier. State machine  220  persistently searches for FB when an FB search window is opened and this machine is selected. 
     The parameters Thresh, Minrise, Max_Fall_Time, and MinFall are programmable by MCU  125 . For those circumstances when CW detection is not required, FB location state machine  220  commences searching for FB at state  000  of  FIG. 4 . If the Ratio&lt;Thresh, then flow continues to state  001 , namely the FB_WAIT_FOR_THRESH state where the state machine waits until Ratio exceeds or equals Thresh. When the Ratio is found to exceed or equal Thresh, then flow continues to state  101 , namely the FB_WAIT_FOR_FALL state. In state  101 , after a Ratio peak has occurred and within the Max_Fall_Time count, Ratio has to fall for no less than the Minfall amount in order to claim that a GSM FB has been detected. Max_fall_time refers to the number of wait counts after a Ratio peak has occurred before going to FB_FRAGMENT_PTM STATE ( 110 ) from the FB_WAIT_FOR_FALL state ( 101 ) if Ratio has not fallen for more than the MinFall amount. If Ratio_fall&gt;Minfall and fb_cnt&lt;=max_fall_time, then an FB is detected and flow continues to state ( 100 ), namely the FB_DETECTED state. Otherwise, flow continues to the FB_FRAGMENT PTM state ( 110 ). Rise_cnt is an internal counter, which counts the number of Ratio[i] occurred while the state machine is in the FB_START_PTM state ( 010 ) and the FB_FRAGMENT_PTM state ( 110 ). Rise_cnt is reset to 0 when the state machine enters the FB_START_PTM state ( 110 ). The state machine  220  will stay in the FB_FRAGMENT_PTM state ( 110 ) until Rise_cnt reaches 15, and afterwards waits until Ratio drops below Thresh. After Rise_cnt reaches 15, once the Ratio falls below Thresh, the state machine returns from the FB_FRAGMENT_PTM state ( 110 ) to the FB_WAIT_FOR_THRESH state ( 001 ). 
     Returning to the FB_SEARCHING state  000  of  FIG. 4 , if Ratio&gt;=Thresh, then flow continues to state  010 , namely the FB_START_PTM state ( 010 ). The FB_START_PTM state  010  uses the MinRise parameter to determine if the ACOR Ratio rises sufficiently fast to be an FB candidate. Ratio_rise is the difference between the current Ratio and the Ratio when the state machine flows to FB_START_PTM state ( 010 ). Ratio_rise provides an indication of the speed at which the Ratio value is increasing. If Ratio_Rise&gt;MinRise then process flow continues to FB_WAIT_FOR_FALL  101  wherein processing continues as described above. However, if Ratio_rise&lt;=MinRise and Rise_cnt=0, the state machine stays at FB_START_PTM state ( 010 ) thus waiting for one more slow-rising or decreasing Ratio[i]. If the next Ratio does not rise high enough or a decreasing Ratio is found, then flow proceeds to the FB_FRAGMENT_PTM state ( 110 ). MCU  125  can instruct either FB location state machine  215  or  220  to be selected by setting an appropriate configuration bit via mode control register  244 . 
     Returning to  FIG. 1 , once an FB is located by either FB location state machine  215  or  220 , frequency offset estimation is commenced. This information is used to change the frequency on which DXCO  130  is operating. As described above, automatic frequency control (AFC) unit  225  in MCU  125  indirectly informs DXCO  130  of the frequency to which DCXO  130  should be tuned so that the offset is ultimately corrected. It is noted that in one particular embodiment, the DCXO operates at approximately 26 MHz whereas the radio channels exist between approximately 850 MHz and approximately 1.9 GHz. The received tone or signal is checked against the expected tone of the FB and, based upon the difference between the two, the DCXO frequency is modified to adjust the 26 MHz locally generated clock signal to a clock recovered from the received tone offset from the expected tone. In actual practice the AFC line represents additional circuitry to assist in retrieving the offset from DSP  200  and communicating the value that the tone is off to the DCXO. Once the offset correction is applied, the local carrier frequency of receiver  120  is tuned to the carrier frequency of the GSM base station. In actual practice it may take several FBs to get sufficiently close to read the SB depending upon the accuracy of crystal  135  and other factors such as the nonlinear characteristics of the DCXO. 
     Once the FB is detected by FB search hardware circuit  185 , FB search hardware circuit  185  may be inactivated to conserve power. Moreover, once FB is detected and after the RF circuitry  115  is deactivated or disabled, then DSP  200  and MCU  125  may be activated. Time domain isolation and noise avoidance is thus achieved. With DSP  200  now active, DSP  200  executes a frequency offset estimation process  205  to estimate the current frequency offset of the received signal. Frequency offset estimation process  205  includes a filter bank process  205 A and a single filter process  205 B. When the FB is initially acquired, the frequency offset can be relatively large and in this case DSP  200  employs the filter bank of filter bank process  205 A. For later received FBs after initial offset correction, the single filter of single filter process  205 B may be used. The operation of filter bank  205 A and filter  205 B by DSP  200  are examples of noise producing digital processes that could potentially interfere with signal reception if they had been allowed to occur while RF circuitry  115  was activated. However in an embodiment discussed above, noise producing digital processes such as those carried out by MCU  125  and DSP  200  were inactivated during RF activity. 
       FIG. 5  is a flowchart illustrating the operation of filter bank process  205 A which is executed while DSP  200  and MCU  125  are active. This is an example of one signal processing activity that may generate digital noise. In one embodiment, during initial FB acquisition, DSP  200  downloads 110 samples of the FB signal from circular buffer  182  of RF receiver interface  180  as per block  500 . The I and Q signals from RF receiver interface  180  are supplied to DSP  200  for this purpose. DSP  200  then computes power (in dB) of the last 100 samples downloaded as per block  505 . After power computation, DSP  200  determines the periodogram of the FB signal using the full 110 sample group as per block  510 . DSP  200  then determines the frequency band showing maximum power in the periodogram as per block  515 . Subsequently in block  520 , DSP  200  sends the full sample set, namely 110 samples in this particular example, through a narrow band filter which covers the frequency band exhibiting maximum power as determined in block  515 . In one embodiment the effective bandwidth of this filter is approximately 20 KHz. DSP  200  then discards the first 10 samples of the narrow band filter output as per block  525 . DSP  200  then computes the power (in dB) of the remaining 100 filter output samples as per block  530  and computes phase of the remaining 100 filter output samples as per block  535 . Then, as per step  540 , DSP  200  computes the phase change of every two successive samples and unwraps the phase change. DSP  200  then applies linear regression to the 100 unwrapped phase changes to provide a frequency offset estimate as per block  545 . Then as per block  550 , DSP  200  determines a quality indicator, QUALITY_INDICATOR, of how accurate the frequency offset estimate is by subtracting the power determined in block  530  from the power determined in block  505 . A quality test is then conducted by the MCU as per decision block  555 . More specifically, if the QUALITY_INDICATOR is not &gt;= a predetermined QUALITY_THRESHOLD, then the frequency offset estimate is discarded and a new FB search is conducted as per block  560 . However, if the QUALITY_INDICATOR is &gt;= to the predetermined QUALITY_THRESHOLD, then the current frequency offset estimate is used to correct the operating frequency of the system as per block  560 . More specifically, the frequency offset estimate is provided to MCU  125  in which AFC unit  225  then instructs DCXO  130  to change its frequency according to a frequency offset correction signal provided on the AFC line coupled to output  225 A. It is noted that, in one embodiment, after RF circuitry  115  is inactivated, DSP  200  is activated to determine the frequency offset estimate as described above. 
       FIG. 6  is a flow chart showing a single filter process  205 B used by DSP  200  once frequency acquisition is achieved and the offset is smaller than during initial acquisition. This is an example of another signal processing activity that may generate digital noise. In one embodiment, DSP  200  downloads 110 samples of the FB signal from circular buffer  182  of RF receiver interface  180  as per block  600 . DSP  200  then computes power (in dB) of the last 100 samples downloaded as per block  605 . In this particular embodiment, a periodogram of the FB is not employed. Rather, DSP  200  sends the full sample set of 110 samples through a narrow band filter as per block  610 . In this embodiment, the center frequency of the narrow band filter is 67.7 KHz. The bandwidth and shape of the filter is the same as the narrow band filter in the process of  FIG. 5 . After this narrow band filtration, in block  615  DSP  200  discards the first 10 samples outputted by the single narrow band filter implemented by DSP  200 . Then DSP  200  computes the power (in dB) of the remaining 100 filter output samples as per block  620  and computes the phase of the remaining 100 filter output samples as per block  625 . DSP  200  now computes the phase change of every two successive samples and unwraps the phase change as per step  630 . DSP  200  then applies linear regression to the 100 unwrapped phase changes to provide a frequency offset estimate as per block  635 . Then, as per block  640 , DSP  200  determines a quality indicator, QUALITY_INDICATOR, of how accurate the frequency offset estimate is by subtracting the power determined in block  620  from the power determined in block  605 . A quality test is then conducted in the MCU as per decision block  645 . More specifically, if the QUALITY_INDICATOR is not &gt;= to a predetermined QUALITY_THRESHOLD, then the frequency offset estimate is discarded and a new FB search is conducted as per block  650 . However, if the QUALITY_INDICATOR is &gt;= to the predetermined QUALITY_THRESHOLD, then the current frequency offset estimate is used to correct the operating frequency of the system as per block  655 . More specifically, the frequency offset estimate is provided to MCU  125  in which AFC unit  225  then instructs DCXO  130  to change its frequency according to a frequency offset correction signal provided on the AFC line coupled to output  225 A. It is noted that, in one embodiment, after RF circuitry  115  is inactivated, DSP  200  is activated to determine the frequency offset estimate as described above. 
     Returning to  FIG. 1 , scheduling of the FB search window is provided by an FB scheduling unit  235  that is implemented as part of the synchronization state machine  230  in MCU  125  in this particular embodiment. The FB search is performed during a search window the timing of which is controlled by MCU  125  through system timer unit (STU)  240  FB scheduling unit  235  of MCU  125  controls the FB search window by setting the time that the FB search is started and stopped via an FB_Search_Enable signal on FB_Search_Enable line  240 B of STU  240 . STU  240  also controls the times at which receiver  120  is activated and inactivated via a POWER DOWN signal on the STU&#39;s POWER DOWN line  240 A. 
     More particularly, the FB search is performed within a search window caused by the STU raising a level sensitive enable signal (the FB_Search_Enable) which is supplied by STU  240  to the FB search hardware (FBSHW)  185 . Upon finding an FB or a CW signal, the FBSHW  185  stops itself even before the window is finished. The FBSHW sets an FB_DONE flag as soon as FB or CW is detected. CW here is used in the sense of being not only a continuous wave signal, but other narrow band interferers as well. If no FB or CW is detected, the FB_DONE flag is set at the end of the search window. The FB_DONE flag is reset at the beginning of each new FB search window. The FB_DONE flag is sent to STU  240  and to circular buffer  182  (connection not shown). STU  240  can choose to use this flag to end the search window immediately. When an FB is detected, STU  240  uses the rising edge of the FB_DONE flag to capture the time value which is then reported back to MCU  125 . In one embodiment, the reported time value refers to the number of quarter bits from the time at which a last instruction issues until the FB_DONE is asserted. Receiver interface  180  then freezes the contents of FB circular buffer  182  such as its write address and stops sending data strobes to the FB search hardware  185 . Once FB or CW is detected by the FB search hardware  185 , an oldest sample pointer (fbEndPointer) is provided by RF RX interface  180 . FB search hardware  185  provides an fb_cnt_value (frequency burst count value) stored in FB status register  222 . When the DSP starts its FB related tasks, it computes a read pointer based on fbEndPointer and fb_cnt_value and provides the read pointer to the circular buffer  182  pointing to the starting position of samples that will be used for frequency offset determination. 
     FB acquisition is performed in a time window, also referred to above as the search window, during which the RF front end namely receiver  120  is continuously active to receive a signal. RF activities such as just described are isolated in time from the baseband digital activities such as those performed by DSP  200  and MCU  125 . This desirably reduces noise in receiver  120 . 
       FIGS. 7A and 7B  together form a flow chart summarizing the operation of wireless communication system  100 . This flow chart is enhanced to show selected signals and subsystems in addition to process flow. The functional blocks of the flowchart are further enhanced to show the active/inactive state of RF receiver (RF RX)  120 , the active/inactive state of the RF RX interface (RF RX INT)  180 , the active/inactive state of frequency burst search hardware (FBSHW)  185  as well as the active/inactive state of DSP  200  and MCU  125 . By “active/inactive state” is meant the active or inactive state of the particular component that is referenced. The active/inactive state or status of referenced components appears in an ellipse adjacent each functional block of the flowchart. 
     Before discussing the flowchart in detail, it is noted that in one embodiment, RF RX interface  180  and FBSHW  185  may remain active or enabled all the time that system  100  is in operation. RF circuitry such as RF RX  120  is cycled through active and inactive status regularly to save power. In yet another embodiment, the FBSHW  185  may be cycled to an inactive state to conserve power when the FBSHW is not being used to locate the FB. In one embodiment, DSP  200  and MCU  125  are active at substantially the same time and inactive at substantially the same time. However generally, RF circuitry such as RF RX  120 , may be activated to enable reception of FB and may be inactivated when or after DSP  200  and MCU  125  are activated to engage in digital processing activities that produce noise. It is noted that in one embodiment, FBSHW  185  is active prior to enabling or activating the RF circuitry since MCU  125  does the enabling. Generally, RF RX interface  180  remains active. 
     In  FIG. 7A , process flow starts when radio frequency (RF) signals are captured by antenna  105  and provided to SAW filter  110  which passes a band of signals as per block  700 . At this point in time, RF circuitry such as RF RX  120  is active. The RF RX interface  180  and FBSHW  185  are also active. However, at this time when the RF circuitry is active, DSP  200  and MCU  125  are held inactive to avoid noise from these devices during reception of RF signals. 
     The RF signal is down-converted to an intermediate frequency (IF) by analog mixer  150  as per block  705 . In one embodiment, DCXO  130  receives a DCXO adjustment value or frequency offset from AFC unit  225  of MCU  125  as per block  710 . In actual practice, MCU  125  sends frequency correction information to an MCU peripheral device (not shown) which passes the information when needed to DCXO  130 . This value is used to adjust the frequency of the DCXO output which is used as an input to frequency synthesizer  140 . In this manner the receive frequency, which is controlled by synthesizer  140 , is indirectly adjusted by changing the value provided to the DCXO. The down-converted analog signal is then converted to a digital signal by analog to digital converters  155  and  160  as per block  715  to provide digital I and Q signals. Digital mixer  165  then down-converts the digital I and Q signal components to baseband as per block  720 . If this is the initial acquisition of FB for a received signal, then the digital signal is supplied to a narrowband FB filter  175 A as per block  725 . However, if FB has already been acquired, then the received signal is supplied to narrowband channelization filter  175 B. As discussed above, the bandwidth of FB filter  175 A is selected to be narrower than the bandwidth of channelization filter  175 B. Channelization filter  175 B is used to receive signals other than signals received during acquisition of FB, i.e. to signals received during normal communication such as for voice and data. FB filter  175 A is used for acquisition of FB and not for normal communication. Dashed line  730  represents a demarcation between RF circuitry activities carried out by transceiver  115  above line  730  and FB search hardware processes and baseband processes which are below line  730 . During the RF circuitry activities described above, the RF circuitry is active while digital circuitry such as MCU  125  and DSP  200  are held inactive to reduce noise. It is noted that above line  730 ,  FIG. 7A  is a data flow diagram whereas below line  730  some paths are displaced in time, i.e. occur after some amount of time has transpired, as described below. Paths  732  and path  742  are examples of such paths displaced in time from the immediately prior step in the flowchart. In path  732  the I and Q samples are passed to FB circular buffer  182  where they can reside for some amount of time before being passed to frequency estimation process  750  which is shown in  FIG. 7B . 
     The received I and Q samples are provided to ACOR computation unit  195  via the FB circular buffer  182 . ACOR computation unit  195  determines the autocorrelation function of the received samples as per block  735 . FBSHW  185  and its ACOR unit  195  are active at this time. The resultant sequence of ratios or values, Ratio[i], are supplied to the frequency burst FB location state machine  210  so that state machine  210  can determine if an FB has been found as well as the location of the FB. Location state machine  210  can search for the FB in two different modes, namely a mode with CW detection and a mode without CW detection as per block  740 . MCU  125  pre-programs the FB location state machine  210  with an FB search mode selection bit in mode control register  244  to instruct the location state machine to use either the mode with CW detection or the mode without CW detection to locate the FB. The FB detection result is then returned to the MCU with FB timing information after the RF circuitry is deactivated. 
     In  FIG. 7B , which shows the remainder of the flowchart, dashed line  745  is used to provide demarcation between FB search hardware (FBSHW) processes above and digital processes of the DSP and MCU below. It is noted that both the DSP and MCU execute code in the performance of their processes. In one embodiment, after the search window expires or after the FB is located, and after receiver  120  is inactivated, DSP  200  and MCU  125  may be activated to commence digital operations. Since receiver  120  is inactive, it is not impaired by the noise generated when DSP  200  and MCU  125  conduct their digital processing tasks. In one alternative embodiment, when the FBSHW locates the FB it inactivates itself automatically, even before the search window finishes, to conserve power. However, prior to going inactive, the FBSHW passes FB location information to a register where the information is stored for later retrieval by DSP  200  and MCU  125  when they become active. In other embodiments, the FBSHW remains active. As seen in the flowchart as per block  750 , the received I and Q samples are provided via FB circular buffer  182  of RF RX interface  180  to the frequency offset estimation process  205  of DSP  200 . The FB detection result and FB timing information are also provided to frequency offset estimation process  205 . MCU  125  instructs frequency offset estimation process  205  whether to use a filter bank or a single filter in determining the frequency offset as per block  750 . The resultant frequency offset estimate is provided to synchronization state machine  230  of MCU  125 . During the above described frequency offset determination processes conducted by DSP  200 , both DSP  200  and MCU  125  remain activated. 
     Dashed line  755  is used to provide demarcation between frequency offset estimation processes conducted by the DSP above and MCU processes below. In this particular instance, line  755  represents a time demarcation as well as a data interface. The FB detection result and FB timing information are provided by the FB search hardware  185  to FB scheduling unit  235  in MCU  125  as per block  760 . The FB scheduling unit  235  instructs STU  240  to tell FB search hardware  185  when to process I and Q samples, i.e. when to open and close the search window as per block  760 . MCU  125  loads this information into STU  240  which controls FBSHW timing. In one embodiment, MCU  125  is inactive or disabled while FBSHW  185  is processing data. FB scheduling unit  235  also instructs FB hardware  185  regarding the FB search mode selection, namely whether to conduct a search with CW detection or without CW detection. DSP  200  provides the AFC unit  225  of MCU  125  with a frequency offset estimate of the current FB also as per block  760 . AFC unit  225  instructs DSP  200  whether to use a filter bank  205 A mode or single filter  205 B mode. In other words, the DSP is provided with a frequency estimation mode selection by AFC unit  225 . AFC unit  225  also provides a DCXO adjustment value to DCXO  130  to correct the frequencies at which synthesizer  140 , and ultimately receiver  120 , are operating. During the times when the above-described digital activities of MCU  125  and DSP  200  are being conducted, the RF circuitry is inactive to save power. It is noted that the RF circuitry can be effectively inactive if it is powered up but signals received by the RF circuitry are ignored. 
     A wireless communication system is thus provided that includes a dedicated FB search hardware (FBSHW) circuit  185  that searches for FB in real time when the RF receiver circuits are active and signal processing circuitry such as the DSP and MCU are inactive. Since the DSP and MCU are inactive during FB acquisition, the noise from digital processes is limited from interfering with reception and acquisition of the FB. The FB search hardware (FBSHW) circuit is dedicated hardware which is designed for low noise operation in comparison with the relatively high amounts of noise typically produced by digital circuits such as the DSP and MCU. In one embodiment, once the dedicated FB search hardware circuit finds the FB and after the RF circuitry has been inactivated, the digital circuitry can be again enabled. Noise problems are thus desirably reduced. 
     Modifications and alternative embodiments of this invention will be apparent to those skilled in the art in view of this description of the invention. Accordingly, this description teaches those skilled in the art the manner of carrying out the invention and is intended to be construed as illustrative only. While GSM embodiments have been described, those skilled in the art will appreciate that the methods taught herein can be applied to other systems as well. The forms of the invention shown and described constitute the present embodiments. Persons skilled in the art may make various changes in the shape, size and arrangement of parts. For example, persons skilled in the art may substitute equivalent elements for the elements illustrated and described here. Moreover, persons skilled in the art after having the benefit of this description of the invention may use certain features of the invention independently of the use of other features, without departing from the scope of the invention.