Patent Publication Number: US-11652447-B2

Title: Power amplifier

Description:
RELATED APPLICATIONS 
     The present application claims the benefit of the filing date of U.S. Provisional Patent Application Ser. No. 62/886,117 which was filed Aug. 13, 2019 and which is hereby expressly incorporated by reference in its entirety. 
    
    
     FIELD 
     The present application relates to power amplifiers and, more particularly, to radio frequency (RF) power amplifiers which are well suited for supporting both low and high power modes of operation and/or which can be implemented in a space efficient manner using field effect transistors. 
     BACKGROUND 
     Antenna arrays to support beam forming often need to be able to vary the power output to perform beam steering and/or other array dependent operations. In such applications the load corresponding to a particular antenna or set of antenna elements may be intentionally controlled to achieve the desired beam shape. As a result, as a beam is steered, the load corresponding to a particular antenna element or set of antenna elements may change, e.g., as part of the beam steering operation. 
     There is a need for power amplifiers which can support a varying loads at frequencies which are likely to be used for radio transmissions, e.g., in GHz frequency ranges and/or other frequency ranges. 
       FIG.  1    is a drawing  100  of showing one known dual input power amplifier  100  including a first power amplifier  102  and a second power amplifier  104 . The dual input power amplifier uses two transistors, transistor 1 as part of the first power amplifier  102  and transistor 2 as part of the second power amplifier, to tailor performance. The dual input power amplifier  100  further includes a combiner  106  to combine the outputs from the two power amplifiers  102 ,  104 . The specific definition of the combiner  106  determines how the individual power amplifiers ( 102 ,  104 ) of the dual input power amplifier  100  operate under back-off output power. Proper design of function [Zcmb], corresponding to the combiner  106 , as a function of not only frequency but power levels results in high efficiency over the output power range. Specific consideration of silicon area to implement [Zcmb] is required for economic advantage. It is desirable to minimize the silicon area needed to implement the combiner, i.e., it is desirable to implement a combiner having a small footprint. It is also desirable for a combiner to operate with high efficiency and high linearity. 
       FIG.  2    is a drawing  200  including the general dual input power amplifier  100  of  FIG.  1    and further including equations  202  defining combiner current and voltage. 
       FIG.  3    is a drawing  300  including the general dual input power amplifier  100  of  FIG.  1    and further includes identifying information defining symbols shown within the main and auxiliary amplifiers ( 102 ,  104 ). β1  302  is the gain of the main amplifier  102 , as indicated by box  304 . β2  308  is the gain of the auxiliary amplifier  104 , as indicated by box  310 . Φ  312  is the phase shift of the auxiliary amplifier  104  relative to the main amplifier  102 . Combiner  106  is a power combiner, as indicated by box  306 . Mathematical approaches treat the values (β1, β2, and Φ) as constants. 
       FIG.  4    is a drawing  400  including the general dual input power amplifier  100  of  FIG.  1    and further identifies that a particular combiner design  404  of the combiner  106  should be based on the device technology  402  being used in the main and auxiliary amplifiers ( 102 ,  104 ). There is a need for determining a “real” implementation of Zcmb which gives best performance (in terms of efficiency) for a given device technology. In particular there is a need for new methods and apparatus for dual input power amplifiers including a combiner implementation which gives best performance (in terms of efficiency) for Complementary Metal Oxide Semiconductor (CMOS) technology, particularly in the microwave frequency ranges. 
       FIG.  5    is a drawing  500  including the general dual input power amplifier  100  of  FIG.  1    and further identifies two prior art load modulated power amplifiers, which may be the dual input power amplifier  100 . The first type of prior art load modulated power amplifier is a Doherty PA  502  and the second type of prior art load modulated power amplifier is a Chireix PA  504 . 
       FIG.  6    is a drawing  600  which is a more detailed representation of the Doherty power amplifier  502  of  FIG.  5   . Doherty power amplifier  502  includes a combiner  604  which includes quarter wave transmission line  602  to combine power and realize load modulation. 
       FIG.  7    is a drawing  700  which is a more detailed representation of the Chireix power amplifier  504  of  FIG.  5   . Chireix power amplifier  504  includes a combiner which includes two quarter wave transmission lines  704 ,  706  to combine power and realize load modulation. 
     Earlier work in the field of Load Modulated Power Amplifiers, e.g., a Doherty PA, uses quarter-wave transmission lines to combine power and realize load modulation. This approach is not feasible with a CMOS integrated solution. In addition to the problem of size, e.g. the transmission lines are relatively large and would take up an unacceptable amount of real estate on the circuit board, there are several other problems (gain compression, phase change across power, and power added efficiency (PAE) under backoff conditions) to consider when using CMOS technology in a dual input power amplifier. 
       FIG.  8    is a drawing  800  including the general dual input power amplifier  100  of  FIG.  1    and describes in box  802  a first load modulated power amp issue: the gain at high power drops near compression due to auxiliary power amp gain (β 2 )  308 . 
       FIG.  9    is a drawing  900  including the general dual input power amplifier  100  of  FIG.  1    and describes in box  902  a second load modulated power amp issue: phase adjustment is needed to match current at the output as the auxiliary PA turns on due to behavior of the PA.  FIG.  10    is a drawing  1000  including the general dual input power amplifier  100  of  FIG.  1    and describes in box  1006  a third load modulated power amp issue: main and auxiliary impedances, represented by arrows  1102 ,  1104 , change with power level to fix power-added efficiency. 
     A drawback of using numerous transmission lines to implement a power amplifier is the space such transmission lines occupy. It would be desirable if in at least some embodiments the use of transmission lines could be reduced or avoided as compared to the known systems. 
     Based on the above discussion, there is a need for new methods and apparatus for power amplifiers which are well suited for the transmission frequency ranges that are likely to be used, e.g., GHz frequency ranges and that such power amplifiers be capable of being implemented in a space efficient manner and/or capable of providing good power efficiency over a range of amplifier power ranges. 
     SUMMARY 
     Methods and apparatus for implementing a power efficient amplifier device through the use of a main (primary) and auxiliary (secondary) power amplifier are described. The primary and secondary amplifiers operate as current sources providing current to the load. Capacitance coupling is used to couple the primary and secondary amplifier outputs. In some embodiments the combination of primary and secondary amplifiers achieve high average efficiency over the operating range of the device in which the primary and secondary amplifiers are used in combination as an amplifier device. The amplifier device is well suited for implementation using CMOS technology, e.g., N-MOSFETs, and can be implemented in an integrated circuit space efficient manner that is well suited for supporting RF transmissions in the GHz frequency range, e.g., 30 GHz frequency range. The primary amplifier in some embodiments is a CLASS-AB or B amplifier and the secondary amplifier is a CLASS-C amplifier. 
     Transmission lines as part of the power amplifier are avoided in some but not necessarily all embodiments. While a transmission line is used at an input to one of the amplifiers in some embodiments transmission lines as part of the output of the power amplifier device are avoided. The amplifiers are well suited in transmission arrays where the load of the amplifier corresponds to an antenna element or set of antenna elements and beam steering may be implemented. 
     An exemplary dual input power amplifier, in accordance with some embodiments of the present invention, includes a combiner with high efficiency and improved linearity, over existing approaches, and can be implemented in a small footprint in comparison to existing approaches which rely on long transmission lines to combine power and realize load modulation. 
     The values of: i) the gain of the main power amplifier (β1), ii) the gain of the auxiliary power amplifier (β1) and iii) the phase of the auxiliary power amplifier (Φ) are not in fact constant, but in reality are actually changing with power which is very important to get a performance improvement in a Complementary Metal Oxide Semiconductor (CMOS) Monolithic Microwave Integrated Circuit (MMIC). 
     While various features discussed in the summary are used in some embodiments it should be appreciated that not all features are required or necessary for all embodiments and the mention of features in the summary should in no way be interpreted as implying that the feature is necessary or critical for all embodiments. 
     Numerous additional features and embodiments are discussed in the detailed description which follows. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         FIG.  1    is a drawing of a general dual input power amplifier including a first power amplifier, a second power amplifier and a combiner. 
         FIG.  2    is a drawing including the general dual input power amplifier of  FIG.  1    and further including equations defining combiner current and voltage. 
         FIG.  3    is a drawing including the general dual input power amplifier of  FIG.  1    and further includes identifying information defining symbols shown within the main and auxiliary amplifiers. 
         FIG.  4    is a drawing including the general dual input power amplifier of  FIG.  1    and further identifies that a particular combiner design of the combiner should be based on the device technology being used in the main and auxiliary amplifiers. 
         FIG.  5    is a drawing including the general dual input power amplifier of  FIG.  1    and further identifies two prior art load modulated power amplifiers, which may be the dual input power amplifier. 
         FIG.  6    is a drawing which is a more detailed representation of the Doherty power amplifier of Figure. 
         FIG.  7    is a drawing which is a more detailed representation of the Chireix power amplifier of  FIG.  5   . 
         FIG.  8    is a drawing including the general dual input power amplifier of  FIG.  1    and describes a first load modulated power amp issue that needs to be taken into consideration: the gain at high power drops near compression due to auxiliary power amp gain. 
         FIG.  9    is a drawing including the general dual input power amplifier of  FIG.  1    and describes a second load modulated power amp issue that needs to be taken into consideration: phase adjustment is needed to match current at the output as the auxiliary PA turns on due to behavior of the PA. 
         FIG.  10    is a drawing including the general dual input power amplifier  100  of  FIG.  1    and describes a third load modulated power amp issue that needs to be taken into consideration: main and auxiliary impedances, represented change with power level to fix power-added efficiency. 
         FIG.  11    is a drawing illustrating a main amplifier and an auxiliary amplifier, each operating as a current source, with their outputs coupled directly to load RL. 
         FIG.  12    is a drawing of an exemplary power amplification device in accordance with an exemplary embodiment, which includes a primary power amplifier (main power amplifier), which is a class AB or class B amplifier, a secondary power amplifier (auxiliary power amplifier), which is a class C amplifier, and a representative combining circuit including a series capacitor CM, coupled together. 
         FIG.  13    is a drawing illustrating a exemplary embodiment for the representative combining circuit of  FIG.  12    in accordance with an exemplary embodiment. 
         FIG.  14    is a drawing illustrating an exemplary power amplification device, in accordance with an exemplary embodiment, including a primary (main) power amplifier, a secondary power amplifier, and a combiner including capacitor CM, wherein the main power amplifier includes stacked MOS FETs to support larger voltages at the load. 
         FIG.  15    illustrates a design method for high power operation (both main and aux power amps on) with regard to an exemplary dual input power amplifier device in accordance with an exemplary embodiment. 
         FIG.  16    illustrates a design method for low power operation (main power amp on and aux power amp off) with regard to an exemplary dual input power amplifier device in accordance with an exemplary embodiment. 
         FIG.  17    illustrates an example for a MNOS Main Amplifier an includes a first Powered Added Efficiency (PAE) contour graphs for 0 dBm input power and a second PAE contour graph for 5 dBm input power. 
         FIG.  18    is a drawing illustrating an example for a MNOS Auxiliary Amplifier and includes a first Powered Added Efficiency (PAE) contour graph for 0 dBm input power and a second PAE contour graph for 5 dBm input power. 
         FIG.  19    is a drawing which illustrates exemplary simulation inputs for exemplary operation of a dual input power amplifier device, which works successfully at 30 GHz, implemented in accordance with an exemplary embodiment. 
         FIG.  20    is a PAE contour graph which illustrates simulation results for exemplary operation of a dual input power amplifier device (corresponding to the exemplary input of  FIG.  19   ) implemented in accordance with an exemplary embodiment. 
         FIG.  21    shows a first exemplary amplifier implemented in accordance with one embodiment of the invention. 
         FIG.  22    shows a second exemplary amplifier implemented in accordance with the invention. 
         FIG.  23    shows a third exemplary amplifier implemented in accordance with the invention. 
         FIG.  24    shows a fourth exemplary amplifier implemented in accordance with the invention. 
         FIG.  25    shows a fifth exemplary amplifier in accordance with the invention. 
         FIG.  26    shows a sixth exemplary amplifier implemented in accordance with the invention. 
     
    
    
     DETAILED DESCRIPTION 
     In accordance with a feature of various embodiments of the present invention, both main and auxiliary amplifiers of a dual input power amplifier operate as current sources directly at the load and achieve high average efficiency. This approach of having both the main and auxiliary amplifiers of a dual input power amplifier operate as current sources directly at the load is not the Doherty approach, which uses long transmission lines to combine power. 
       FIG.  11    is a drawing  1100  illustrating a main amplifier  1102  and an auxiliary amplifier  1104 , each operating as a current source, with their outputs ( 1103 ,  1105 ) coupled directly to load RL  1106 . The output impedance of the amplifiers is modulated by the auxiliary current. 
     The output impedance (ZM) of the main amplifier  1102  is modulated by the current (iA) of the auxiliary amplifier  1104 , as indicated by box  1108 . The output impedance (ZA) of the auxiliary amplifier  1104  is modulated by the current (iM) of the main amplifier  1102 , as indicated by box  1110 . 
     One disadvantage of this approach is that the main amplifier  1102  now sees a large voltage swing. In accordance with a feature of various embodiments of the current invention, a “stacked” power amplifier is implemented, which can support the larger voltage at load. For example, the main power amplifier  1102  includes a plurality of NMOS FETS, e.g. 4 NMOS FETs, in a stacked configuration and the auxiliary power amplifier  1104  includes a plurality of NMOS FETs, e.g. 4 NMOS FETs, in a stacked configuration. 
     Various aspects and/or features of power combining, in accordance with various embodiments of the present invention will some be described.  FIG.  12    is a drawing of an exemplary power amplification device  1200  in accordance with an exemplary embodiment. Power amplification device  1200  includes a primary power amplifier (main power amplifier)  1202 , which is a class AB or class B amplifier, a secondary power amplifier (auxiliary power amplifier)  1204 , which is a class C amplifier, and a combiner  1206  coupled together as shown. The power combining approach of device  1200  is current combining, which deviates from the Doherty design approach of using a quarter wave transformer for combining. 
       FIG.  13    is a drawing  1300  illustrating an representation of a model (shown for AC operation) for an exemplary combiner  1206 , in accordance with an exemplary embodiment, coupled to a transistor  1302  of the primary (main) amplifier  1202 , and a transistor  1304  of the secondary (auxiliary) amplifier  1204 . The model for the combiner includes an inductor LM  1306 , which is the inductor connected to the power supply for the primary (main) power amp  1202 , an an inductor LA  1308 , which is the inductor connected to the power supply for the secondary (auxiliary) power amp  1204 , a capacitance Co,,M  1310 , which is an output capacitance of the primary (main) power amplifier  1202 , a capacitance Co,,A  1312 , which is an output capacitance of the secondary (auxiliary) power amplifier  1204 , a load RL  1314 , a capacitor CM  1316 , and a capacitor CDC, BLOCK  1318 , coupled together as shown. 
     Series capacitor CM  1316  is chosen based on asymmetric PA bias (bias differences between primary PA  1202  and secondary power amp  1204 ) and auxiliary power amp capacitance variation (Co,,A  1312  capacitance variation, e.g., between low power mode and high power mode). Co,A  1312  and Co,M  1310  are “inherent” output device capacitances and change with output power in CMOS process. Co,A  1312  changes with power due to class-C biasing. 
       FIG.  14    is a drawing illustrating an exemplary power amplification device  1400  including a primary (main) power amplifier  1402 , a secondary power amplifier  1404 , and a combiner  1406 . The primary power amplifier  1402  includes 4 N MOS FETs ( 1408 ,  1410 ,  1412 ,  1414 ) coupled together in a stacked configuration, with the bottom FET  1408  receiving input signal RFIN 1   1416  on its gate, and which each of the other FETs ( 1410 ,  1412 ,  1414 ) gate inputs being coupled to ground via a capacitor ( 1418 ,  1420 ,  1422 ), respectively. The secondary power amplifier  1404  includes 4 N MOS FETs ( 1424 ,  1426 ,  1428 ,  1430 ) coupled together in a stacked configuration, with the bottom FET  1424  receiving input signal RFIN 2   1432  on its gate, and which each of the other FETs ( 1426 ,  1428 ,  1430 ) gate inputs being coupled to ground via a capacitor ( 1434 ,  1436 ,  1438 ), respectively. 
     Exemplary combiner  1406 , is coupled to the drain of transistor  1414  of the primary (main) amplifier  1402 , and to the drain of transistor  1430  of the secondary (auxiliary) amplifier  1404 . The model for the combiner includes an inductor LM  1440 , which is the inductor connected to the power supply VDD  1442  for the primary (main) power amp  1402 , an inductor LA  1444 , which is the inductor connected to the power supply VDD  1446  for the secondary (auxiliary) power amp  1404 , a capacitance Co,,M  1448 , which is an output capacitance of the primary (main) power amplifier  1402 , a capacitance Co,,A  1450 , which is an output capacitance of the secondary (auxiliary) power amplifier  1404 , a load RL  1452 , a capacitor CM  1454 , and a capacitor CDC, BLOCK  1456 , coupled together as shown. 
       FIG.  15    is a drawing  1500  including a drawing illustrating a representation of a model (shown for AC operation as inductors are shown connected to gnd) for an exemplary combiner  1501 , in accordance with an exemplary embodiment, coupled to a transistor  1502  (e.g., drain of N MOS FET) of the primary (main) amplifier, and a transistor  1504  (e.g., drain of N MOS FET) of the secondary (auxiliary) amplifier. The model for the combiner  1501  includes an inductor LM  1506 , which is the inductor connected to the power supply for the primary (main) power amp, an inductor LA  1508 , which is the inductor connected to the power supply for the secondary (auxiliary) power amp, a capacitance Co,,M  1510 , which is an output capacitance of the primary (main) power amplifier, a capacitance Co,,A  1512 , which is an output capacitance of the secondary (auxiliary) power amplifier, a load RL  1514 , a capacitor CM  1516  (used for load modulation of the main power amp), and a DC blocking capacitor CDC, BLOCK  1518 , coupled together as shown. 
     Series capacitor CM  1516  is chosen based on asymmetric PA bias (bias differences between primary PA and secondary power amp) and auxiliary power amp capacitance variation (Co,,A  1512  capacitance variation, e.g., between low power mode and high power mode). Co,A  1512  and Co,M  1510  are “inherent” output device capacitances and change with output power in CMOS process. Co,A  1512  changes with power due to class-C biasing. 
     Co,M  1510  is substantially the same for high power and low power operation. Co,A  1512  is significantly different for high power and low power operation, as indicated by the representation of Co,A as a variable capacitor with the arrow. This is shown in the NMOS Main and Auxiliary Amplifier simulation shown in  FIGS.  17  and  18   .
         Drawing  1500  also includes drawing  1503 , which is a high power representation of the exemplary combiner  1501 . Co,A (HP)  1512 ′ is the high power representation of Co,A  1512 .       

     LM  1506  is defined to be nearly resonant with Co,M  1510 , as indicated by oval  1518 , where the remaining reactance from resonating out CoM  1510  is no more than 40% of the original LM and is referred to as LM,eff  1522 . 
     LA  1508  is defined to be nearly resonant with Co,A (HP)  1512 ′, as indicated by oval  1520 , where, the remaining reactance from resonating out CoA(HP)  1512 ′ is no more than 40% of the original LA. 
     Drawing  1500  also includes drawing  1505 , which is a High-Power Equivalent model. LM,eff  1522  is a small amount of residual inductance, which remains after most of the inductance of LM  1506  is absorbed by the resonance with Co,M  1510 . LM, eff  1522  is typically less than 40% of LM  1506 . In some embodiments, LM, eff  1522  is less than 10% of LM  1506 . 
     The combination of LA  1508  and Co,A(LP)  1512 ″ (low power output capacitance of auxiliary amp—see  FIG.  16   ) produces an effective inductance LA,eff  1524 . The auxiliary capacitor Co,A  1512  is changing with power. Co,A (HP)  1512 ′ is more than 200% higher than Co,A(LP)  1512 ″, and ΔCo,A  1526  is used to indicate the portion of CoA (HP)  1512 ′ corresponding to the change in power, e.g., from low power mode (aux amp off) to high power (aux amp On). 
     The CoA  1512  at high power, represented by Co,A (HP)  1512 ′ ( FIG.  15   ) is typically more than 200% higher than it was at low power represented by Co,A(LP)  1512 ″ (see  FIG.  16   ). 
     At ω0 inductances from load inductors (LM  1506  and LA  1508 ) on the main and auxiliary PAs resonate with output capacitance Co,M  1510  and CO,A(HP)  1512 ′ leaving similar loadline matching at both the main and auxiliary ports ( 1528 ,  1530 ) to the load RL  1514 . RHP is approximately 2RL. RHP is the “loadline” resistance that both the main and auxiliary amplifiers want to see at high power. A 50 ohm load (RL) would be matched with a 100 ohm resistance from the main PA and a 100 ohm resistance from the auxiliary PA. 
     CM  1516  adds some series impedance that is absorbed into the power matching while operating in the high power mode. 
       FIG.  16    is a drawing  1600  including a drawing illustrating a representation of a model (shown for AC operation as inductors are shown connected to gnd) for an exemplary combiner  1501 , in accordance with an exemplary embodiment, coupled to a transistor  1502  (e.g., drain of N MOS FET) of the primary (main) amplifier, and a transistor  1504  (e.g., drain of N MOS FET) of the secondary (auxiliary) amplifier. Co,M is substantially the same for high power and low power operation. Co,A is significantly different for high power and low power operation, as indicated by the representation of Co,A as a variable capacitor with the arrow. Combiner representation  1501  has been previously described with respect to  FIG.  15   . 
     Low power mode will now be described. Auxiliary PA is off. Output capacitance CoA drops when compared to high power operation, thus CoA(LP)  1512 ″&lt;CoA(HP)  1512 ′. (Co,A (HP)  1512 ′ is typically more than 200% higher than Co,A(LP)  1512 ″). The combination of LA  1508  and Co,A(LP)  1512 ″ produces an effective inductance LA,eff  1524  that produces an impedance transformation network to reduce the load impedance at the main PA. CM  1516  and Leff, M  1522  and Leff, A  1524  resonate at ω to change the impedance seen at the main amplifier. 
     Drawing  1600  also includes drawing  1602 , which is a low power representation of the exemplary combiner  1501 . Co,A (LP)  1512 ″ is the low power representation of Co,A  1512 . LM  1506  is defined to be nearly resonant with Co,M  1510 , as indicated by oval  1604 , where the remaining reactance from resonating out CoM  1510  is no more than 40% of the original LM and is referred to as LM,eff  1522 . 
     Drawing  1600  also includes drawing  1603 , which is a Low-Power Equivalent model. LM,eff  1522  is a small amount of residual inductance, which remains after most of the inductance of LM  1506  is absorbed by the resonance with Co,M  1510 . LM, eff  1522  is typically less than 40% of LM  1506 . In some embodiments, LM, eff  1522  is less than 10% of LM  1506 . LA,eff  1524  is an effective inductance which is the combination of LA  1508  and Co,A(LP)  1512 ″. 
     C M    1516  and Leff, M  1522  and Leff, A  1524  resonate at ω0, as indicated by oval  1610 , to change the impedance seen at the main amplifier. 
     RLP is approximately 4RL. The RLP is the “loadline” resistance that the main amplifier want to see at low power at the main port  1528 . A 50 ohm load (RL) would be matched with a 200 ohm resistance from the main PA. At the auxiliary port  1530 , RLP is large, e.g., more than at least eight times RL. 
       FIG.  17    is a drawing  1700  illustrating an example for a NMOS Main Amplifier. Drawing  1700  includes a first Powered Added Efficiency (PAE) contour graph  1702  for 0 dBm input power and a second PAE contour graph  1704  for 5 dBm input power. 
     Under normal circumstances, the impedance of the main PA changes from A ( 1706 )→B ( 1708 ), which is a change from 100 ohms to 50 ohms, with increasing power level. This is not useful for load modulation. 
     For load modulation, it is desirable for the impedance to change from C ( 1710 )→D ( 1712 ) with increasing power level. 
     A Doherty approach would use a transmission line to implement C→D. But if considering the loss from a quarter transmission line (TL) of 0.7-1 dB in Doherty, no improvement would be found by using the Doherty approach. 
     In accordance with an approach using an embodiment of the present invention, one would match device at high power to 100 ohms and design the combining cap C M  such that the impedance is transformed to 50 ohms at low power. 
       FIG.  18    is a drawing  1800  illustrating an example for a NMOS Auxiliary Amplifier. Drawing  1800  includes a first Powered Added Efficiency (PAE) contour graph  1802  for 0 dBm input power and a second PAE contour graph  1804  for 5 dBm input power. 
     The auxiliary power amplifier is biased in class-C and indicates a high loadline impedance and capacitance for low input power, as indicated by the location of dot  1806  on the first PA PAE contour graph  1804 . 
       FIG.  19    is a drawing  1900  which illustrates exemplary simulation inputs for exemplary operation of a dual input power amplifier device implemented which works at 30 GHz in accordance with an exemplary embodiment. Set of information  1902  identifies the start, stop and step frequencies (1 GHz, 50 GHz, and 100 MHz) of the simulation. Set of information  1904  identifies values of Lm,eff, La,eff, Cm, and Cs used, which work at 30 GHz. Circuit  1906  is a low power model of the circuit with values of Lm,eff, La,eff and Cm that map to an implementation that works at 30 GHz. Circuit  1908  is a high power model of the circuit with values of Lm,eff, La, eff, Cm, and Cs (additional capacitance added by auxiliary amp as it turns on) that map to an implementation that works at 30 GHz. It may be observed that the load resistance ( 1910 ) for the low power model ( 1906 ) is 50 ohms, and that the load line resistance Z ( 1912 ) is 200 ohms. (RP=4RL) It may be observed that the load resistance ( 1914 ) for the high power model ( 1914 ) is 50 ohms, and that the load line resistance Z ( 1916 ) is 100 ohms. (RP=2RL). 
       FIG.  20    is a PAE contour graph  2000  which illustrates simulation results for exemplary operation of a dual input power amplifier device (corresponding to the exemplary input of  FIG.  19   ) implemented in accordance with an exemplary embodiment. Block  2002  indicates that in low power, the auxiliary amp is off, and in high power the auxiliary amp reaches high power. Dot A  2006  on graph  2006  indicates high power. Dot B  2004  in graph  2005  indicates low power. 
     Various aspects and/or features of the design method will now be discussed. The CM series capacitor, used in various embodiments in accordance with the present invention, addresses a design technique to implement a power combiner with only a capacitor that introduces far less loss and area requirements than a transmission line. Capacitors in CMOS processes have significantly higher Q than transmission lines reducing losses. This characteristic gives the approach, in accordance with the present invention, of using the CM series capacitor for combining, a significant advantage over Doherty approaches for load modulation. 
     Solutions for other design issue problems will now be discussed. A first issue of concern is gain compression. An auxiliary amplifier (biased in class-C) exhibits a gain variation that does not compensate for the main amplifier gain compression. A solution to this problem, in accordance with a feature of some embodiments of the present invention includes designing, implementing and using a neutralization network in the class-C amplifier (not connected to the output). The neutralization network is designed to increase the gain at high power levels relative to low power levels. 
     A second issue of concern is phase variation. In accordance with a feature of some embodiment of the present invention, the power amplifier is implemented as a “dual input” power amplifier such that one can use phase shifters to dynamically change the phase into the two paths of the power amplifier. 
     A real CMOS PA has several issues. It should be able to handle high voltage swings. In accordance with a feature of some embodiments of the present invention, stacked FET PAs are used in the implementation. It is desirable to eliminate feedback through the PA. In accordance with a feature of some embodiments of the present invention, utilization network(s) are used. It is desirable to improve backoff efficiency. In accordance with a feature of some embodiments of the present invention, the implementation allows supply modulation. 
     The stacked FET increases the loadline impedance to be a current source. The main amplifier is assumed to be biased in class AB or class B, while the auxiliary amplifier is biased in class C. In put power is split between the PAs (main and aux), while the auxiliary amplifier is twice the size of the main amplifier. 
     A triangle is used to illustrate the power amplifiers in  FIGS.  21  to  26    but due to size constraints the stack of transistors and capacitors which form the amplifier are not necessarily all shown inside the triangle representation the individual power amplifier. Accordingly, simply because a component is not inside a triangle does not mean it is not part of the primary or secondary power amplifier in the corresponding figure and elements which are described as being part of a particular amplifier should be considered a component of the amplifier whether it is shown internal or external to a triangle representing a particular amplifier to which the component corresponds. 
       FIG.  21    shows a first exemplary amplifier device  2100  implemented in accordance with one embodiment of the invention that uses a primary power amplifier  2201  in combination with a secondary power amplifier  2208 . The primary and secondary power amplifiers are arranged in parallel with the output of both amplifiers  2201  and  2208  being on during high power mode of device operation. During the low power mode of operation the secondary amplifier  2208  is off but the primary power amplifier  2201  is on. Regardless of the mode of operation DC power, e.g., the gate bias voltages, are supplied to both amplifiers during both the low and high power modes of operation. 
     As explained above the capacitance of the secondary power amplifier  2208  changes significantly, e.g., more than doubles, when the secondary power amplifier  2208  transitions from the off/low power mode of amplifier operation to the high power mode of operation. The impendence and capacitance of the primary amplifier  2201  remains relatively constant in both the low and high power modes of operation since the primary amplifier  2201  remains on in both the low and high power modes of operation. 
     The power amplifiers of  FIGS.  22 - 26    support low and high power modes of operation where which are the same or similar to that described with regard to the  FIG.  21    embodiment. Thus during high power mode of operation the secondary amplifiers shown in  FIGS.  22 - 26    will be on but off during low power mode of operation. Similar changes in the capacitance of the secondary power amplifiers occurs in the devices shown in  FIGS.  22 - 26    with the capacitance of the secondary amplifier at least doubling and often more than doubling when the secondary amplifier changes from the off/low power mode of device operation to the high power mode of device operation in which the secondary amplifier is on and contributes current to the output load. 
     The amplifier devices shown in  FIGS.  21  to  26    are well suited for RF transmission applications, e.g., to support RF radio signal transmissions via a transmitter array. In some such embodiments the load RL corresponds to an antenna element or set of antenna elements of the antenna array being used to support radio transmissions. The amplifier device shown in  FIGS.  21 - 26    can and in some embodiments is used to support device embodiments where beam steering is implemented. 
     In the  FIG.  21    embodiment, the amplifier  2100  operates as a current-combined, dual input PA with the primary, e.g., main amplifier  2201  operating as a class-AB or class-B amplifier and the secondary (auxiliary) power amplifier  2208  operating as a Class-C power amplifier. The bias voltage source  2514  is shown as a Class B source but would be a Class-AB bias voltage source in the case where the primary amplifier is operated as a Class-AB amplifier. Separate RF signal inputs  2206 ,  2236  for the primary  2201  and secondary  2208  power amplifier are provided in the  FIG.  21    embodiment. In particular the primary amplifier includes a first RF signal input RFin, 1   2206  while the secondary power amplifier  2208  includes a second RF signal input RFin, 2   2236 . 
     In various embodiments the RF signals supplied to the primary and secondary amplifiers are different in at least one of phase and/or amplitude. In some embodiments the signal supplied to the secondary power amplifier  2208  is derived from the signal supplied to the first RF signal amplifier  2201 . As will be discussed below in some embodiments a network using inductors and capacitors is used to generate the signal supplied to the second amplifier  2208  by applying an intentional phase shift into the signal supplied to the first RF signal amplifier  2201  to generate the signal supplied to the secondary amplifier  2208 . Such an approach avoids the use of transmission lines as part of the power amplifier device. In other embodiments a circuit including a transmission line is used to generate the signal supplied to the second amplifier  2208  by introducing an intentional phase shift into the signal supplied to the first signal amplifier  2201  to thereby generate the second amplifier signal input. By introducing a phase shift between signals RFin, 1  and RFin, 2  it is possible to prevent the output signals for the primary and secondary amplifiers being out of phase. Such a phase difference might occur between the amplifier output signals if one or more of the amplifier introduced a phase shift that was not taken into consideration and compensated for at the signal input to the auxiliary amplifier. 
     The power amplification device  2100  includes a primary power amplifier  2201  including one or more field effect transistors (FETs)  2202 ,  2204  arranged in a first stack  2233 . The gate  2240  of a first FET  2204  is coupled to a CLASS-B or CLASS AB DC bias voltage source  2514  with a DC blocking capacitor  2211  being used to couple the gate  2240  to the first RF signal input  2206 . The gate of the second FET  2204  is coupled to grand by a capacitor  2271 . The output of the primary amplifier SO 1   2213  is coupled to a first voltage source VDD  2209  by a first inductor  2208  and is also coupled to the output load RL  2220 . 
     The secondary power amplifier  2208  includes a second stack  2233  of FETs  2232 ,  2234  with the gate  2260  of the first FET  2234  in the second stack  2233  being coupled to the second F input  2236  via a second DC blocking capacitor  2241 . The gate of the second capacitor  2232  in the second stack  2233  is coupled to ground by capacitor  2270 . The drain  2238  of the secondary power amplifier  2208  serves as the output  2238  of the second power amplifier  2208  and is coupled to the load  2220 . 
     Thus from  FIG.  21    it can be seen that the device  2100  includes two amplifiers  2201 ,  2208  which are arranged in parallel but which receive different RF signal inputs and are biased differently, e.g., with the first amplifier being biased to operate as a Class-AB or Class-B amplifier and the secondary amplifier  2208  being biased to operate as a Class-C amplifier. 
       FIG.  22    shows an amplifier device  2300  which is similar to the amplifier  2100  but with a capacitor CM  2311  being used to couple the output of the primary amplifier  2301  to the output of the secondary amplifier  2238  and load RL  2505 . The components of the amplifier device  2200  are coupled together as shown in  FIG.  22   . Series capacitor CM is chosen in some embodiments based on the asymmetric power amplification bias voltages used and the secondary power amplifier capacitance variation between the on high capacitance during the “on” high power mode of operation and lower capacitance of the secondary amplifier  2331  during the low power “off” mode of operation. CM  2311  adds some series impedance that is absorbed into the power matching. CM series capacitance allows the power combining to be implemented with a capacitor and without the need for a transmission line. This approach introduces far less loss and area requirements than a transmission line. Furthermore in CMOS implementations, e.g., implementations using N-MOSFETs, this is particularly useful since capacitors in CMOS processes have significantly higher Q than transmission lines reducing losses as compared to embodiments where transmission lines are used on the output for power combining. 
       FIGS.  23  through  26    show various exemplary amplifier implementations. The same reference numbers are used in these figures to refer to the same or similar elements. Accordingly, once an element is described it may not be described again in detail in the later embodiment in which the same reference number is used. 
     In various ones of the  FIG.  23 - 26    embodiments stacked FET power amplifiers are used to support high voltage swings. In at least some embodiments. Feedback through the power amplifiers is reduced or eliminated through the use of a neutralization network sometimes implemented as a unilateralization network. The methods allow supply modulation and can improve backoff efficiency as compared to other designs. The stacked FETs in the power amplifiers naturally increase the loadline impedance to be a current source. Input power is split between the power amplifiers in the  FIGS.  22 - 26    embodiments while the secondary amplifier is twice the size of the primary amplifier or even larger in some embodiments. 
       FIG.  23    shows a fourth exemplary amplifier  2300  implemented in accordance with the invention. The exemplary amplifier  2300  is similar to the one shown in  FIG.  22    but includes a larger stack of transistors, e.g., 4 transistors, in each of the amplifiers  2501 ,  2531  and also includes in each of the primary and secondary amplifiers  2501 ,  2531  a unilateralization network  2545 . In the case of the primary amplifier  2501 , the unilateralization network  2545  is used and in the case of the secondary amplifier  2531  the unilateralization network  2575  is used. 
     The first stack of FETS  2503  included in the primary amplifier  2501  includes a first N-MOSFET  2516 , a second N-MOSFET  2514 , a third N-MOSFET  2508  and a fourth N-MOSFET  2504 . The gates of the second, third and fourth N-MOSFETS are connected to ground through capacitors  2543 ,  2542  and  2540  respectively. The gate  2540  of the first N-MOSFET  2516  is coupled to the CLASS-B DC bias voltage source  2514  and is also coupled via DC blocking capacitor  2513  to the first RF signal input  2506 . The drain of the last N-MOSFET  2504  in the first stack  2503  of FETs serves as the first signal output SO 1   2513  and is coupled, via coupling capacitor CM  2511 , to the load RL  2520  and further via DC blocking capacitor  2521  to the second signal output SO 2   2538  of the secondary amplifier  2531 . Series capacitor CM  2511  between the primary and secondary power amplifiers  2501 ,  2531  is used to realize a load modulation of the main amplifier impedance and in various embodiments is less than ⅓ the size of the DC blocking capacitor  2521  but in many embodiments less than ⅕ the size of the DC blocking capacitor  2521 . In some embodiments the DC blocking capacitors  2521 ,  2513  and  2541  are each at least 3 times the size of CM  2511  but in some cases at least 5 times the size of CM  2511 . In this way the DC blocking capacitors will block DC but pass the RF signal being amplified. 
     The secondary amplifier  2531  includes a stack of at least 4 N-MOSFETs  2554 ,  2552 ,  2534  and  2532 . The second stack of FETs  2533  included in the secondary amplifier  2531  includes a first N-MOSFET  2554 , a second N-MOSFET  2552 , a third N-MOSFET  2534  and a fourth N-MOSFET  2532 . The gates of the second, third, and fourth N-MOSFETS are connected to ground through capacitors  2573 ,  2572  and  2570 , respectively. The gate of the first N-MOSFET  2554  is coupled to the CLASS-C DC bias voltage source  2544  and is also coupled via DC blocking capacitor  2541  to the second RF signal input  2536 . Thus the second RF signal input is protected from the DC bias voltage but allows the second RF signal supplied to RF signal input  2536  to pass. The drain of the last N-MOSFET  2532  in the second stack  2533  of FETs is coupled to the second signal output SO 2   2538 . 
     The first signal output is coupled SO 1   2513  is coupled to the voltage source VDD,M by the first inductor  2511  while the second signal output SO 2  is coupled to voltage source  2539  VDD,A by second inductor  2527 . The RF output signal current output by the primary and secondary amplifiers  2501 ,  2531  is supplied to load RL  2520  from the first and second outputs SO 1 , SO 2  via capacitor CM  2511  and the blocking capacitor  2521 , respectively with the separate DC voltage sources VDD,M  2509  and VDD,A  2539  being separated through the use of the capacitors. 
     Note that in the  FIG.  23    embodiment the first unilateralization network  2545  spans across multiple transistors  2508 ,  2514  and extends from the drain of the third transistor  2505  to the gate  2540  of the first transistor  2516  in the first stack  2503  of transistors. Similarly the second unilateralization network  2575  spans across multiple transistors  2552 ,  2534  and extends from the drain  2564  of the third transistor  2532  to the gate  2560  of the first transistor  2554  in the second stack  2533  of transistors. 
       FIG.  24    shows an amplifier device  2500  which is the same as the one shown in  FIG.  23    but with more detail provided with regard to the neutralization networks  2545 ,  2575 . In the  FIG.  24    exampled each of the first  2545  and second  2575  neutralization networks are implemented using a DC blocking capacitor in combination with an inductor. In particular in  FIG.  24    the first neutralization network  2545  includes DC blocking capacitor  2546  and third inductor  2548 . The second neutralization network  2575  includes DC blocking capacitor  2576  and fourth inductor  2578 . 
       FIG.  25    shows an amplifier device  2588  which is the same as the one shown in  FIG.  24    but with a π network  2592  coupling the first RF signal input  2506  to the second RF signal input  2536 . The π network  2592  includes two inductors  2596 ,  2594  connected to ground. The two inductors  2596  and  2594  are further coupled together by a capacitor  2592 . The π network  2592  is used to apply a phase shift to the RFin, 1  to generate the RFin, 2 . The introduced phase shift compensates for a phase shift introduced by one or both of the primary and secondary amplifiers so that the signals output by the two amplifiers are synchronized. The use of the  7 C network avoids the need for transmission lines to introduce phase or signal delays and is particularly well suited for CMOS implementations since it avoids the use of relatively large transmission lines to introduce phase or signal delay. Thus the  FIG.  25    is particularly well suited for compact implementations requiring relatively small amounts of chip area as compared to embodiments where transmission lines are used as part of the amplifier device. The  FIG.  25    embodiment which may be described as a current-combined, dual input power amplifier can address phase offset for large-signal compression mismatch through the use of the 3-port network  2590 . 
       FIG.  26    shows an amplifier device  2600  which is similar to the one shown in  FIG.  25    but in which a component  2600  including a transmission line is used to implement a phase shift. In the  FIG.  26    network the transmission line component  2602  coupled the first RF signal input  2506  to the second RF signal input  2536  with the transmission line in component  2602  introducing a desired phase shift and/or signal delay into the first RF signal to produce the second RF input signal. As with the  FIG.  25    embodiment the output of the primary and secondary amplifiers  2501  and  2531  will be synchronized. While the  FIG.  26    embodiment includes a transmission line between the two amplifier inputs, it avoids the lossy and lengthy TL at the output of some of the known power amplifiers discussed in the background section of the application. 
     For some simulated embodiments 36% peak PAE and 27.9% 6-dB PAE are achieved in simulation results but there is no guarantee that all embodiments will produce such results.
         The neutralization network is implemented in some embodiments as a unilaterization. The neutralization network is used in some embodiments to improve gain as compared to embodiments where such a network is not used. The number of transistors in each of the stacks can be increased and the number of transistors in each stack is exemplary and not intended to be limiting.       

     Numbered List of Exemplary Apparatus Embodiments 
     Apparatus Embodiment 1 A power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) comprising: a primary power amplifier ( 2201  or  2501 ) including one or more field effect transistors (FETs) ( 2202 ,  2204  or  2504 ,  2508 ,  2514 ,  2516 ), said primary power amplifier ( 2201  or  2501 ) being a class AB power amplifier or a class-B power amplifier and having a first signal input ( 2206  or  2506 ) for receiving a first input signal (RFin, 1 ) and a first signal output (SO 1 ) ( 2213  or  2513 ) coupled to a load (RL) ( 2220  or  2520 ); and a secondary power amplifier ( 2208  or  2531 ) including one or more FETs ( 2232 ,  2234  or  2532 ,  2534 ,  2552 ,  2554 ), said secondary power amplifier ( 2208  or  2531 ) being a class-C power amplifier having a second signal input ( 2236  or  2536 ) for receiving a second input signal (RFin, 2 ), a second signal output (SO 2 ) ( 2238  or  2538 ) coupled to the load ( 2220  or  2520 ), said first signal output (SO 1 ) ( 2213  or  2513 ) and said second signal output (SO 2 ) ( 2238  or  2538 ) being coupled together. 
     Apparatus Embodiment 2 The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 1, wherein both said primary power amplifier ( 2201  or  2501 ) and secondary power amplifier ( 2208  or  2531 ) are coupled to a first DC voltage source (VDD) ( 2209  or  2509 ) by a first inductor (LM) ( 2208  or  2507 ) which is coupled to said first (SO 1 ) signal output ( 2213  or  2513 ) and said second ( 2238  or  2538 ) signal output (SO 2 ). 
     Apparatus Embodiment 3 The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 2, wherein the primary ( 2201  or  2501 ) power amplifier includes at least two N-MOSFETs ( 2202 ,  2204  or  2504 ,  2508 ,  2514 ,  2516 ) arranged in a first stack ( 2203  or  2503 ) and the secondary power amplifier ( 2208  or  2531 ) also includes at least two N-MOSFETs ( 2232 ,  2234  or  2532 ,  2534 ,  2552 ,  2554 )) arranged in a second stack ( 2233  or  2533 ). 
     Apparatus Embodiment 3A The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 3, wherein arranging the N-MOSFETs of the first power amplifier ( 2201  or  2501 ) in the first stack ( 2203  or  2503 ) combines power while increasing a loadline resistance of the first power amplifier ( 2201  or  2501 ) as compared to a case where fewer N-MOSFETs are used in the first stack ( 2203  or  2503 ). 
     Apparatus Embodiment 3B The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 3A, wherein arranging the N-MOSFETs of the second power amplifier ( 2208  or  2531 ) in a second stack ( 2233  or  2533 ) combines power while increasing the loadline resistance of the second power amplifier ( 2208  or  2531 ) as compared to a case where fewer N-MOSFETs were used in the second stack ( 2233  or  2533 ). 
     Apparatus Embodiment 4 The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 2, wherein the first signal input ( 2206  or  2506 ) receives a first RF signal and where the second signal input ( 2236  or  2536 ) receives a second RF signal, said second RF signal being different from said first RF signal in at least one of phase or amplitude. 
     Apparatus Embodiment 4.01 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 1, wherein said first signal output (SO 1 ) ( 2513 ) of the primary power amplifier ( 2501 ) is coupled to a first DC voltage source (VDD, M) ( 2509 ) by a first inductor (LM) ( 2507 ); and wherein the first signal output (SO 1 ) ( 2513 ) is coupled to the load ( 2520 ) by a first coupling capacitor CM ( 2511 ). 
     Apparatus Embodiment 4A The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4, wherein the first coupling capacitor CM ( 2511 ) is the simplest embodiment of a network that provides load modulation to the main amplifier ( 2501 ). 
     Apparatus Embodiment 4AA The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4, wherein the secondary power amplifier ( 2531 ) is off when the power amplification device is operating in a low power mode of operation and on when operating in a high power mode of operation, the capacitance of the secondary power amplifier ( 2531 ) increasing by at least a factor of two when the secondary power amplifier turns on (note that in the case of being on or off a DC bias gate voltage is still supplied to one or more gates to FETS included in the secondary amplifier). 
     Apparatus Embodiment 4AAA The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4AA, wherein the capacitance of the primary power amplifier ( 2501 ) remains substantially the same (e.g., remains within 10% of its maximum capacitance) when operating in both the low and high power modes of operation. 
     Apparatus Embodiment 4AAC The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4AA, wherein during the high power mode of operation the capacitance of the secondary amplifier ( 2531 ) and inductance of the second inductor (LA) ( 2537 ) substantially cancel each other out (e.g., resonate within 20% of the natural frequency of the capacitance and inductance) at the operating frequency ω0. 
     Apparatus Embodiment 4B The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4, wherein the capacitor CM ( 2511 ) in combination with capacitance of the secondary amplifier ( 2531 ) and inductance of the second inductor (LA) ( 2537 ) (capacitance of secondary amp in low power mode in combination with LA—form effective inductance which resonates with CM) resonate at a frequency ω0 when the power amplification device is in a low power mode of operation and the secondary amplifier ( 2531 ) is off, said frequency ω0 being the intended operating frequency of the power amplification device. 
     Apparatus Embodiment 4BB The power amplification device of Apparatus Embodiment 4B, wherein during the low power mode of operation the capacitor CM ( 2511 ) in combination with capacitance of the secondary amplifier ( 2531 ) resonates out at least 80% of the effective inductance of the second inductor (LA) ( 2537 ). 
     Apparatus Embodiment 4C The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4, wherein said coupling capacitor CM ( 2511 ) provides a signal path for modulating the impedance seen by the output of the first power amplifier ( 2501 ) as the impedance of the secondary amplifier ( 2531 ) changes based on the amount of current supplied by the secondary amplifier ( 2531 ). 
     Apparatus Embodiment 5 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 4, wherein said second signal output (SO 2 ) ( 2538 ) of the secondary amplifier ( 2531 ) is coupled to a second supply voltage source (VDD, A) ( 2539 ) by a second inductor (LA) ( 2537 ); and wherein the second signal output (SO 2 ) ( 2538 ) is coupled to the load ( 2520 ) by a first direct current (DC) blocking capacitor CDC,BLOCK ( 2521 ). 
     Apparatus Embodiment 5A The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 5, where the first DC blocking capacitor ( 2521 ) has a capacitance at least 3 times larger (but sometimes 5 time or more larger) than the first coupling capacitor CM ( 2511 ). 
     Apparatus Embodiment 6 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 5, wherein the primary power amplifier ( 2501 ) includes a first stack ( 2503 ) of at least two N-MOSFETs ( 2505 ,  2508 ,  2514 ,  2516 ); wherein a gate ( 2540 ) of a first N-MOSFET ( 2516 ) of the first stack ( 2503 ) of at least two N-MOSFETs ( 2504 ,  2508 ,  2514 ,  2516 ) is coupled by a second DC blocking capacitor ( 2513 ) to the first RF signal input ( 2506 ) and to a DC class-B bias voltage source ( 2514 ). 
     Apparatus Embodiment 7 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 6, wherein the secondary power amplifier ( 2531 ) includes a second stack ( 2533 ) of at least two N-MOSFETS ( 2532 ,  2534 ,  2552 ,  2554 ); and wherein a gate ( 2560 ) of a first N-MOSFET ( 2554 ) of the second stack ( 2533 ) of at least two N-MOSFETS ( 2532 ,  2534 ,  2552 ,  2554 ) is coupled by a third DC blocking capacitor ( 2541 ) to the second RF signal input ( 2536 ) and to a DC CLASS-C bias voltage source ( 2544 ). 
     Apparatus Embodiment 8 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 6 further comprising: a first neutralization network ( 2545 ) coupled between a drain ( 2505 ) of one of the N-MOSFETS ( 2508 ) in the first stack ( 2503 ) of MOSFETS and the gate ( 2540 ) of the first N-MOSFET ( 2516 ) in the first stack ( 2503 ) of MOSFETS. 
     Apparatus Embodiment 8A The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 8, wherein the first neutralization network ( 2545 ) provides a signal feedback path across multiple FETS ( 2508 ,  2514 ) in the first stack ( 2503 ) of MOSFETS. 
     Apparatus Embodiment 8B The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 8A, wherein the first neutralization network ( 2545 ) is a unilateralization network. 
     Apparatus Embodiment 9 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 8, wherein said first neutralization network ( 2545 ) includes a fourth DC blocking capacitor ( 2546 ) and a third inductor ( 2548 ). 
     Apparatus Embodiment 10 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 9 further comprising: a second neutralization network ( 2575 ) coupled between a drain ( 2564 ) of one of the N-MOSFETS ( 2534 ) in the second stack ( 2533 ) of MOSFETS and the gate ( 2560 ) of the first N-MOSFET ( 2554 ) in the second stack ( 2533 ) of MOSFETS. 
     Apparatus Embodiment 10A The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 8, wherein the second neutralization network ( 2575 ) provides a signal feedback path across multiple MOSFETS ( 2534 ,  2552 ) in the second stack ( 2533 ) of MOSFETS. 
     Apparatus Embodiment 10B The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 10A, wherein the second neutralization network ( 2575 ) is designed to peak gain under high-power conditions when the second power amplifier ( 2531 ) is operating in an on state. 
     Apparatus Embodiment 11 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 10, wherein said second neutralization network ( 2575 ) includes a fifth DC blocking capacitor ( 2576 ) and a fourth inductor ( 2578 ). 
     Apparatus Embodiment 12 The power amplification device ( 2588 ) of Apparatus Embodiment 7, further comprising: a capacitor and inductive network ( 2590 ) coupling the first signal input ( 2506 ) to the second signal input ( 2536 ), said capacitor and inductive network ( 2590 ) introducing at least a phase shift into a first RF signal received at the first RF signal input to produce a second RF signal supplied to the second RF signal input ( 2536 ). 
     The introduced phase shift is chosen in at least some embodiment based on a phase shift introduced in the output due to the modulation network (CM). 
     Apparatus Embodiment 12A The power amplification device ( 2588 ) of Apparatus Embodiment 12, wherein capacitor and inductive network ( 2590 ) is a Pi network. 
     Apparatus Embodiment 13 The power amplification device ( 2600 ) of Apparatus Embodiment 12, further comprising: a link ( 2602 ) including a transmission line coupling the first signal input ( 2506 ) to the second signal input ( 2536 ), said link ( 2602 ) introducing at least a phase shift into a first RF signal received at the first RF signal input ( 2506 ) to produce a second RF signal supplied to the second RF signal input ( 2536 ). 
     Apparatus Embodiment 14 The power amplification device of Apparatus Embodiment 1, wherein both the primary ( 2201  or  2501 ) and secondary power amplifier ( 2208  or  2531 ) operate in the same frequency range (e.g., where in some embodiments the frequency range is a 20 to 40 GHz frequency range). 
     Apparatus Embodiment 15 The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 1, wherein during a low power mode of operation, the primary power amplifier ( 2231  or  2531 ) is ON and the secondary power amplifier is OFF (but power is still supplied to the secondary power amplifier ( 2231  or  2531 ), e.g., the secondary power amplifier&#39;s gate are still supplied with the Class-C DC bias voltage but does not supply a significant amount of current to the load). 
     Apparatus Embodiment 16 The power amplification device ( 2100 ,  2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 15, wherein during a high power mode of operation, both the primary power amplifier ( 2201  or  2501 ) is ON and the secondary power amplifier is ON ( 2208  or  2531 ) (e.g., supplying current to the load). 
     Apparatus Embodiment 17 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 16, wherein during the high power mode of operation, LM ( 2507 ) resonates with capacitance of the primary amplifier ( 2501 ) and LA ( 2537 ) resonates with capacitance of the secondary amplifier ( 2531 ). 
     Apparatus Embodiment 18 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 16, wherein during the high power mode of operation, a loadline resistance of the primary power amplifier ( 2501 ) is approximately the same as (e.g., within 20%) the loadline resistance of the secondary power amplifier ( 2531 ). 
     Apparatus Embodiment 19 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 18, wherein the loadline resistance of the primary power amplifier ( 2501 ) during the high power mode of operation is approximately twice the resistance of the load ( 2520 ) (e.g., 2 RL). 
     Apparatus Embodiment 20 The power amplification device ( 2200 ,  2300 ,  2500 ,  2588  or  2600 ) of Apparatus Embodiment 15, wherein during the low power mode of operation, the loadline resistance of the primary power amplifier ( 2501 ) is approximately four times the resistance of the load ( 2520 ) (e.g., 4 RL or approximately 4 RL). 
     Numbered List of Exemplary Method Embodiments 
     Method Embodiment 1 An amplification method, the method comprising: supplying a first radio frequency signal to a first signal input ( 2206  or  2506 ) of a primary power amplifier ( 2201  or  2501 ), the primary power amplifier ( 2201  or  2501 ) including one or more field effect transistors (FETs) ( 2202 ,  2204  or  2504 ,  2508 ,  2514 ,  2516 ), said primary power amplifier ( 2201  or  2501 ) being a class AB power amplifier or a class-B power amplifier and having the first signal input ( 2206  or  2506 ) for receiving a first input signal (RFin, 1 ) and a first signal output (SO 1 ) ( 2213  or  2513 ) coupled to a load (RL) ( 2220  or  2520 ); and supplying a second radio frequency signal to a second signal input ( 2236  or  2536 ) of a secondary power amplifier ( 2208  or  2531 ), the secondary power amplifier ( 2208  or  2531 ) including one or more FETs ( 2232 ,  2234  or  2532 ,  2534 ,  2552 ,  2554 ), said secondary power amplifier ( 2208  or  2531 ) being a class-C power amplifier having a second signal input ( 2236  or  2536 ) for receiving a second input signal (RFin, 2 ), a second signal output (SO 2 ) ( 2238  or  2538 ) coupled to the load ( 2220  or  2520 ), said first signal output (SO 1 ) ( 2213  or  2513 ) and said second signal output (SO 2 ) ( 2238  or  2538 ) being coupled together. 
     Method Embodiment 2 The method of Method Embodiment 1, further comprising: performing at least a phase shifting operation on the first radio signal input to generate said second radio frequency signal from said first radio frequency signal. 
     It is understood that the specific order or hierarchy of steps in the processes and methods disclosed is an example of exemplary approaches. Based upon design preferences, it is understood that the specific order or hierarchy of steps in the processes and methods may be rearranged while remaining within the scope of the present disclosure. The accompanying method claims present elements of the various steps in a sample order, and are not meant to be limited to the specific order or hierarchy presented. In some embodiments, one or more processors are used to carry out one or more steps of the each of the described methods. 
     In various embodiments each of the steps or elements of a method are implemented using one or more processors. In some embodiments, each of elements or steps are implemented using hardware circuitry. 
     In various embodiments nodes and/or elements described herein are implemented using one or more components to perform the steps corresponding to one or more methods, for example, controlling, establishing, generating a message, message reception, signal processing, sending, communicating, e.g., receiving and transmitting, comparing, making a decision, selecting, making a determination, modifying, controlling determining and/or transmission steps. Thus, in some embodiments various features are implemented using components or in some embodiments logic such as for example logic circuits. Such components may be implemented using software, hardware or a combination of software and hardware. Many of the above described methods or method steps can be implemented using machine executable instructions, such as software, included in a machine readable medium such as a memory device, e.g., RAM, floppy disk, etc. to control a machine, e.g., general purpose computer with or without additional hardware, to implement all or portions of the above described methods, e.g., in one or more nodes. Accordingly, among other things, various embodiments are directed to a machine-readable medium, e.g., a non-transitory computer readable medium, including machine executable instructions for causing a machine, e.g., processor and associated hardware, to perform one or more of the steps of the above-described method(s). Some embodiments are directed to a device, e.g., a SPI interface device, a chip, a device including an array of chips with using a common SPI interface bus, a wireless communications device including a multi-element antenna array supporting beam forming, such as a cellular AP or Wifi AP, a wireless terminal, a UE device, etc., including a processor configured to implement one, multiple or all of the steps of one or more methods of the invention. 
     In some embodiments, the processor or processors, e.g., CPUs, of one or more devices, are configured to perform the steps of the methods described as being performed by the devices, e.g., communication nodes. In some but not all embodiments a device, e.g., a wireless communications node such as an access point, base station or user device, includes one or more amplifier devices implemented in accordance with the invention. The components may be implemented using software and/or hardware. 
     Some embodiments are directed to a computer program product comprising a computer-readable medium, e.g., a non-transitory computer-readable medium, comprising code for causing a computer, or multiple computers, to implement various functions, steps, acts and/or operations, e.g. one or more steps described above. Depending on the embodiment, the computer program product can, and sometimes does, include different code for each step to be performed. Thus, the computer program product may, and sometimes does, include code for each individual step of a method. The code may be in the form of machine, e.g., computer, executable instructions stored on a computer-readable medium, e.g., a non-transitory computer-readable medium, such as a RAM (Random Access Memory), ROM (Read Only Memory) or other type of storage device. In addition to being directed to a computer program product, some embodiments are directed to a processor configured to implement one or more of the various functions, steps, acts and/or operations of one or more methods described above. Accordingly, some embodiments are directed to a processor, e.g., CPU, configured to implement some or all of the steps of the methods described herein. The amplifier device of the invention may be for use in, and part of, e.g., a chip or other circuit and may be and often is used in a wireless communications device. 
     Numerous additional variations on the methods and apparatus of the various embodiments described above will be apparent to those skilled in the art in view of the above description. Such variations are to be considered within the scope. Numerous additional embodiments, within the scope of the present invention, will be apparent to those of ordinary skill in the art in view of the above description and the claims which follow. Such variations are to be considered within the scope of the invention.