Patent Publication Number: US-2010130145-A1

Title: Amplification system for interference suppression in wireless communications

Description:
This application claims the benefit under Title 35, United States Code, Section 119 and incorporates by reference Korean applications 10-2008-0116908, filed Nov. 24, 2008; 10-2008-0116929, filed Nov. 24, 2008; 10-2008-0116958, filed Nov. 24, 2008; 10-2008-0116971, filed Nov. 24, 2008; 10-2008-0118909, filed Nov. 27, 2008; and 10-2008-0118915, filed Nov. 27, 2008. 
     BACKGROUND 
     The present invention generally relates to amplification of wireless signals in communication equipment. More specifically, the present invention relates to amplification of wireless signals efficiently. 
     Mobile telecommunication networks employ stationary communication units such as base stations and repeaters to allow communications between wireless devices, such as cell phones and other devices. The repeaters are used between the base station and the wireless devices to enhance the quality of the RF signal, extend service area around the base stations and reduce the cost of the network. The output power of a base station is as large as several hundred Watts. The average output power of a repeater varies from a few Watts to about sixty Watts. However, the power output efficiency of equipment in stationary communication units is notoriously “low”, at only a little better than ten percent. The output RF power efficiency for the purposes of the present invention is defined as: total RF radiation power of the stationary communication unit divided by DC electric power required by an output power amplifier (PA) of the stationary communication unit in order to generate that total RF radiation power. So for a stationary communication unit having a power output efficiency of ten percent, about 200 Watts of DC power at the output power amplifier is required to radiate a useful 20 Watts RF power signal to the open space through an antenna. The remaining 180 Watts of the 200 Watts of DC power is lost as heat, which should be removed quickly for the stability of the system. To maintain stable equipment operation, the excess heat generated by this loss usually requires a heat sinking passive panel, as well as an active fan and air or water cooling devices to remove the heat from the system. 
       FIG. 1  shows a schematic of a typical RF power amplification system used in current stationary communication units. There is input RF signal to be amplified. The input RF signal is inputted at point (a) to a Driving Amplifier (DA). The output of the DA is a first amplified version of the RF signal that was inputted to the DA. The output of the DA is then inputted to an output power amplifier (PA) at point (c). The output of the PA is a second amplified version of the RF signal that was inputted to the DA and which is outputted to open space at point (d) to via an antenna (ANT). In the current amplification systems of  FIG. 1 , the input RF signal is amplified less at the DA, than at the PA, and the greater amplification is performed at the PA. 
     One of the reasons for such low power efficiency of mobile telecommunication equipment is that the quality of RF signal radiated to an open space needs to be extremely high. This requirement of a high quality signal is necessary for preventing interference among signals from different service providers in common open space, as required by laws in many countries. Among several characteristics in the radiation of a RF, Adjacent Channel Leakage Power Ratio (ACLR) is one of the most important characteristics to be considered to prevent interference among RF signals from different service providers. The optimum efficiency of the PA can be obtained, in general, when the PA is operating at near its saturation point. Most PA exhibit some degree of nonlinearity near their saturation point, which causes an increase in the spectral growth of the output power density and leads to distortion of the ACLR of the output signal. Therefore, current PAs employed in typical amplification systems are designed to operate within a linear region prior to the saturation point of the PA to satisfy the ACLR requirement and therefore sacrifice efficient operation of the PA. 
       FIG. 2  shows a single channel power spectral density graphical representation using a typically system of the class related to current amplification systems that are depicted in  FIG. 1 . Using the properties of isolation and sharp skirt together, the ACLR can be expressed graphically in output signal power spectrum density readouts, as shown in  FIG. 2 .  FIG. 3  shows a power spectral density graphical representation of a full band WIBRO RF signal at point (a) depicted in  FIG. 1 .  FIG. 4  shows a power spectral density graphical representation of the full band WIBRO RF signal of  FIG. 3  at point (c) depicted in  FIG. 1 .  FIG. 5  shows a power spectral density graphical representation of a twenty Watts full band WIBRO output RF signal at point (d) depicted in  FIG. 1 . The ACLR for a twenty Watt full band WIBRO output RF signal of  FIG. 5  shows about −29 dBc, which does not meet the current ACLR requirements of equal or better than −37 dBc. A PA used in the output power amplification systems of  FIG. 1  has to operate at its linear region with the lower output power efficiency in order to meet the required ACLR of −37 dBc or better. Consequently, an output RF signal strength of full band WIBRO of  FIG. 1 , would be much smaller than twenty Watts with much lower output power efficiency. 
     In general, the efficiency of a RF power amplifier transistor is better than twenty five percent. It is reported that an efficiency of even close to fifty percent with a gain of 8 db to 20 db is available for newly developed power amplifier transistors. The quality of the final output signal is not only dependent on the characteristics of PA of  FIG. 1 , but is affected strongly by the quality of the input RF signal to the PA. Whereby, both the in-band RF signal and out-band noise are amplified at PA. Usually the gain of the PA is between 30 dB to 50 dB. This means that a magnitude of 0 dBm input signal is required to generate a 30 to 50 dBm output signal. However, the out-band noise is also amplified by 30 dB to 50 dB to produce out-band noise in the range of −20 dBm or higher, which is not a very desirable situation in terms of maintaining the required ACLR characteristics when producing an amplified RF output signal. 
     If the RF power efficiency of a base station or repeater could be raised from ten percent to twenty percent for an example, the benefits would not only be from the savings of electric energy cost, but also from the savings of manufacturing, installation, maintaining, and durability of the equipment due to their simpler, lighter, and smaller configuration compared to those used in current systems. It is very desirable to enhance the efficiency of generating a high quality useful RF radiation signal in the mobile telecommunication equipment. The mobile WIMAX, WIBRO and fourth generation mobile telecommunication networks, such as the LTE (Long Term Evolution), are planned for 2009 and beyond in the United States, as well as other parts of world. The output power levels for the planned network equipment are quite high, from fifty Watts to a few hundred Watts. It is clear that higher efficiency power equipment will desirable to be employed with the larger output RF power equipment. The demand for high efficient RF output power of the mobile telecommunication equipment for base stations and repeaters is on the increase. It would be a big step forward to improve the efficiency of mobile telecommunication equipment by finding a relatively simple way to enhance the efficiency of the output power amplifier in the stationary comm units to enhance whole networks including the mobile WIMAX, WIBRO and the up coming the fourth generation systems such as the LTE. It is worthwhile try to understand why the efficiency of output power amplifiers in mobile communication equipment is so low. 
     It is an object of the present invention to provide an amplification system for wireless communications that operates near optimum efficiency while suppressing interference. 
     SUMMARY OF THE INVENTION 
     An amplification system including a high gain amplifier, filter module and low gain amplifier. The high gain amplifier for receiving an input RF signal and processing the input RF signal to produce a first amplified signal while the high gain amplifier is operating near its saturation point. The filter module having at least one band pass filter to receive the first amplified signal and process the first amplified signal to remove unwanted characteristics of the first amplified signal to produce a processed first amplified signal. The low gain amplifier receiving the processed first amplified signal and processing the processed first amplified signal to produce a second amplified signal that has an increase in signal strength over the processed first amplified signal while the low gain amplifier is operating near its saturation point. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a schematic view of a typical amplification system in the prior art. 
         FIG. 2  is a graphical data representation of output power spectral density according to the present invention. 
         FIG. 3  is a graphical data representation of a signal at the position (a) of  FIG. 1  according to the present invention. 
         FIG. 4  is a graphical data representation of a signal at the position (c) of  FIG. 1  according to the present invention. 
         FIG. 5  is a graphical data representation of a twenty Watt output signal at the position (d) of  FIG. 1  according to the present invention. 
         FIG. 6  is a schematic view of an amplification system according to the present invention. 
         FIG. 7  is a graphical data representation of a signal at the position (a′) of  FIG. 6  according to the present invention. 
         FIG. 8  is a graphical data representation of a signal at the position (b′) of  FIG. 6  according to the present invention. 
         FIG. 9  is a graphical data representation of a signal at the position (c′) of  FIG. 6  according to the present invention. 
         FIG. 10  is a graphical representation of characteristics of a RF band pass filter according to the present invention. 
         FIG. 11  is a schematic view of two band pass filters connected in series according to the present invention. 
         FIG. 12  is a graphical representation of characteristics of two RF band pass filters connected in series according to the present invention. 
         FIG. 13  is a schematic view of two band pass filters connected in series according to the present invention. 
         FIG. 14  a schematic view of a plurality of band pass filters connected in series according to the present invention. 
         FIG. 15  is a graphical representation of the ideal response according to the present invention. 
         FIG. 16  is a graphical data representation of a  20 W output signal at the position (d′) of  FIG. 6  according to the present invention. 
         FIG. 17  is a table of experimental data according to the present invention. 
         FIG. 18  is a schematic view of WIBRO repeater with the amplification system according to the present invention. 
         FIG. 19  is a representation of principles of pre-distorter linearization according to the present invention. 
         FIG. 20  is a schematic view of DPD according to the present invention. 
         FIG. 21  is a schematic view of DPD with the amplification system according to the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     The present invention is an amplification system to suppress interference in mobile telecommunication equipment, while increasing RF power output efficiency. The present invention is also a method of implementing the suppression of interference in mobile telecommunication equipment, while increasing RF power output efficiency of the in mobile telecommunication equipment and maintaining the required ACLR. Whereby, RF power output efficiency is defined as: total RF radiation power of the stationary communication unit divided by DC electric power required by an output power amplifier of the stationary communication unit in order to generate that total RF radiation power. The amplification system of the present invention provides signal characteristics of a large isolation, sharp skirt, a good ripple, and acceptable S 11  and S 12  properties. 
     The amplification system of the present invention includes a High Gain Driving Amplifier (HGDA), Filter Module (FM), and a Linearization RF Power Amplifier (LA), as shown in  FIG. 6 . In order to evaluate and compare the data of the present invention shown in  FIGS. 6-16  with the data presented for the prior art shown in  FIGS. 1-5 , the targeted output signal power was chosen when using the HGDA-FM-LA combination during experimentation was the same as the output signal power recorded for the DA-PA combination. The input RF signal to be amplified and outputted from a communications unit enters at point (a′) into the HGDA, as depicted in  FIG. 6 .  FIG. 7  shows an example of the input RF signal at point (a′) into the HGDA, as depicted in  FIG. 6 . Notice that the ACLR of  FIG. 3  and  FIG. 7  is about the same value as −32 dB to evaluate and compare the amplification systems of  FIG. 1  and  FIG. 6 . The HGDA is a high gain amplifier. The function of HGDA is to generate a large pre-determined gain to the input RF signal and deliver the amplified RF signal to the FM and the LA. A magnitude of gain in the range of about 60 dB to 80 dB is envisioned at the HGDA, which is much larger than that of the conventional DA depicted in  FIG. 1 . The gain generated in the HGDA while the HGDA is operating at or near its saturation point, in order to provide that the amplifier used as the HGDA is operating at or near optimal efficiency of the amplifier.  FIG. 8  shows a first amplified version of the input RF signal of  FIG. 7 , which is depicted at point (b′) in  FIG. 6 . The HGDA is chosen based on the amplifier&#39;s output level and optimizing the amplifier&#39;s efficiency, and is less concern with its output signal quality, as shown in  FIG. 8 . This is because the input RF signal to the LA will be improved significantly by the FM. The FM includes one or more Band Pass Filters (BPF). The FM can also include additional components to improve the signal processing of the first amplified version of the input RF signal, as will be describe further. The one or more BPF of the FM are used to improve the first amplified version of the input RF signal to meet ACLR requirements. This is shown in  FIG. 9 , which shows the first amplified version of the input RF signal of  FIG. 7  at point (c′) after the signal has been filtered to produce a filtered amplified version of the input RF signal of  FIG. 7 . The FM is setup to produce an extremely clean signal with specific properties, as shown in  FIG. 9  for the LA input. This is because the LA is to be designed to operate at near its saturation point for optimum power output efficiency with the pass-in quality. 
       FIG. 10  depicts an example of the characteristics of a RF band pass filter (BPF).  FIG. 11  is a schematic diagram of two RF band pass filters connected in series. The RF band pass filters described through out the present invention can be of various types, including a metal cavity, dielectric, strip line, elliptic function type, coaxial line, PBAR, and so on. When more than one RF band pass filter is used, there can be a combination of all above different types of RF band pass filters.  FIG. 12  depicts an example of the characteristics of two RF band pass filters of  FIG. 10  connected in series. Notice that the isolation and skirt properties which affect ACLR have been improved twice from −50 dB to −100 dB and from −50 dB/delta f to −100 dB/delta f, respectively. However, an insertion loss and ripple become degraded twice, from −5 dB to −10 dB and from −10 dB to −20 dB, respectively. By connecting several high quality RF band pass filters in series, the ability to obtain larger isolation and skirt values is achieved. For an example, if a number “N” of RF band pass filters is connected in series for the BPF depicted in  FIGS. 10 and 11 , then the final isolation and skirt values will be N×(−50 dB) and “N×(−50 dB/delta f)”, respectively. Insertion loss and ripple will also increase by “N×(−5 dB)” and “N×(−5 dB)”, respectively, for the BPF depicted in  FIGS. 10 and 11 . Insertion loss can be compensated for by installing a Low Gain Linear Amplifier (LGLA) between RF band pass filters, as shown in  FIG. 13 . The LGLA is usually a low gain linear power amplifier used to make up for signal loss during filtering of a signal. 
     A more difficult task is the improvement of the ripple property, as the ripple property deteriorates by connecting several RF BPFs in series. Prevention of ripple property deterioration can be solved by connecting, in series, a ripple compensating circuit (RCC), as depicted in  FIG. 14 . The RCC can be designed by using known band stop or directional filters.  FIG. 15  depicts the BPF characteristics of  FIG. 10  to produce an ideal response with low ripple and insertion loss properties, after the RF signal has been processed through the LGLA, RCC and BPFs depicted in  FIG. 14 . Note, that close to the “ZERO” for the insertion loss and ripple in terms of absolute values, indicates the superior properties of a design combination of LGLA, RCC and BPFs to from the FM. RCCs and LGLAs are used as deem necessary by the designer of the amplification system when designing the FM for specific applications. The RCCs and LGLAs can be removed or reduced by designing or selecting RF BPFs properly. It is desirable to have a tunable impedance matching tunable circuit for coupling between each of the RCC, LGLA and RF BPF connected in series to optimize the coupling between them for the maximum output. The impedance matching tunable circuit between every two components in the FM can be important. Proper impedance matching of components in the FM reduces reflection of the signal when transitioning from one component to another component. Proper impedance matching is also important between the HGDA and FM, as well as between the FM and the LA. 
     The LA is a power amplifier having a gain of not much more than 20 dB to replace the conventional PA and to produce the second amplified version of the input RF signal that will be outputted from the stationary communication unit. The LA is a low gain amplifier. The amplifier used as the LA should be is operating at or near its saturation point when producing the gain in the RF signal, in order to provide that the amplifier used as the LA is operating at or near optimal efficiency of the amplifier.  FIG. 16  shows a 20 Watt second amplified version of the input RF signal of  FIG. 7  at point (d′), after the LA processes the amplified version of the input RF signal of  FIG. 7  from point (c′). The gain of the LA is usually chosen to be less than 15 dB. The LA is designed to operate near its saturated region to optimize an efficiency of the LA. Even though first amplified version of the input RF signal is amplified at a non-linear region of the LA with very high efficiency, the second amplified version of the input RF signal output power density spectrum looks like the signal was amplified linearly as shown in  FIG. 16 . The ACLR for a twenty Watt full band WIBRO output RF signal of  FIG. 16  is shown to be −38 dB, which does meet the current ACLR requirement. The signal looks like the signal was amplified linearly for two reasons. First, since the gain of LA is designed to be not more than 20 dB instead of usual 30 dB to 50 dB of the conventional PA, the noise level amplified by the LA becomes at least 30 dB less than that produced by the conventional PA, thereby providing an output noise level of similar to current conventional amplification systems. Second, the quality of first amplified version of the input RF signal to the LA from FM is much better quality than that to the conventional PA, as shown by the comparison of  FIG. 4  and  FIG. 9 . The ACLR of  FIG. 9  is about
     −30 dB better than that of  FIG. 4 .   

     The table of  FIG. 17  shows real measured data of preample power of a WIBRO full band for two different LA amplifiers. Where LA 1  is a class B amplifier and LA 2  is a class AB amplifier. The FM used for producing the data in the table of  FIG. 17  are made of two different kinds of filters. One is a 12 poles DR cavity filter to minimize the insertion loss and the other is a 14 poles metal cavity filter for filtering out unwanted higher order harmonics. The size of FM is about 211 mm×100 mm×70 mm. The insertion loss and skirt characteristics are −3 dB and −80 dB/0.5 MHz, respectively. One well tuned impedance matching device is used between two filters. Of course the FM can be designed various ways as has been described. The table shows that for both the LA 1  and LA 2 , the ACLR is −37 dB or better, which is acceptable for ACLR requirements. The efficiency of LA 1  and LA 2  using the HGDA and FM combination is better than 40% at a WIBRO full band output power level of 20 Watts.  FIG. 18  shows a block diagram of a WIBRO repeater with the HGDA-FM-LA combination of  FIG. 6  for providing a stationary communication unit having high RF output power efficiency. Antennas (ANT) are shown receiving and transmitting RF signals. An input RF signal from one or two ANT and amplified by an LGLA to an appropriate magnitude to supply an input RF signal to the HGDA is shown. The signal from the S/W LNA is amplified by HGDA to have a predetermined large enough gain in signal strength. This gain at the HGDA is filtered by FM to pass in-band signal and reject out-band noise sufficiently to obtain very a large isolation output signal from the FM. The signal from the FM supplies the LA with a cleaner version of the signal with the predetermined gain to provide for a desired magnitude RF output signal from the LA with satisfactory ACLR, EVM, and other required properties. 
     As a theoretical example, it will be explained how to determine the approximate amount of gain required at each amplifier of the HGDA-FM-LA combination. One of the variables that controls the output strength of the RF signal is gain at the LA, which has been determined to be optimal between 10 and 20 dB. If one desires an output RF signal of 100 Watt from a stationary communication unit, one would require a 50 dbm signal. One might choose an amplifier for the LA that has a 15 dB gain while operating at its saturation point. Therefore the strength of the signal from the FM should be 35 dBm, because 35 dBm plus 15 dB equals 50 dbm. It has been shown in experimentation that a properly designed FM causes a loss of −3 dB in signal strength. Therefore the signal strength should be at 38 dBm prior to entering the FM, in order to have a 35 dBm signal to enter the LA. Next, the strength of the input RF signal and the choice of the HGDA must be coordinated to produce a 38 dBm signal prior to entering the FM. As an example, the combination of an input RF signal of −32 dBm and a HGDA that generates a 70 dB gain while operating at its saturation point would produce a 38 dBm signal. The −32 dBm input RF signal is a signal that has been received and processed by the communication unit for various known reasons to be at −32 dBm. Working backwards in this manner during design produces a more precise amplification system that provides high gains while attempting to prevent self-oscillation due to parasitic feedback at the receiving antenna of the stationary communication unit. 
     The amplification system using the HGDA-FM-LA combination can produce gains in signal strength without sacrificing optimum power output efficiency. This because unlike the conventional systems currently in use, the two amplifiers employed are operating at or near optimal efficiency for each amplifier. The HGDA-FM-LA combination can be applied for the TDD (time division duplex) of WIBRO or mobile WIMAX, FDD (frequency division duplex) of WCDMA and again TDD of the 4 th  generation LTE (Long Term Evolution) systems. In addition to above RF Power output efficiency enhancement by amplification system of the present invention, the HGDA-FM-LA combination also contributes on the Higher Data Rate and Spectral Efficiency, which is the efficiency of data delivery capability of the communication network. For an example, the higher spectral efficiency system requires less RF power output to cover a certain area than for lower efficiency network system. This is because the quality of RF output signal and the capability of cleaning a noisier input signal is provided by using the HGDA-FM-LA combination of the present invention. 
       FIGS. 19 and 20  depict a known method that uses a signal processor referred to as Digital Pre-Distortion (DPD), which is used with the conventional PAs of  FIG. 1 .  FIG. 19  shows the DPD and the components used with the DPD to aid in processing the signal to be strengthened.  FIG. 20  shows the principles of the DPD technique, where combine processing of the signal with the DPD and PA in a non linear state produces an output signal that has properties as if the signal were process by an amplifier that produces gain in a linear fashion. In DPD method, the input RF signal has been converted to a digital form before entering the Crest Factor Reduction unit (CFR), so that the signal may be processed by the DPD. The input RF signal is modified due to signal processing by the DPD engine in real time using the digital form of the input RF signal and using the digitally transformed feedback of the analog output signal from the PA at a coupler in such a way as to correct or improve the ACLR of the output power density spectrum. The signal from the DPD travels through an up converter frequency mixer than to the PA, but the signal must first be converted to analog using a Digital to Analog Converter (DAC). The feedback signal from the output signal of the PA is a small percentage of the output signal from the PA. That small percentage of the output signal from the PA is converted to a digital form by traveling through a down converter frequency mixer. The down converter frequency mixer attached after the ADC is also attached to a Local Oscillator (LO) to cause the down conversion of the frequency. The down converter frequency mixer outputs the converted signal to an Analog to Digital Converter (ADC). The converted digital of the feedback signal from the PA is fed back to the DPD. Note, that in  FIG. 19 , there is an up converter frequency mixer between the DPD and PA that is also attached to the LO. The up converter frequency mixer along with the LO up converts the signal from the DPD after it has left the DAC. The DPD method requires a very fast micro-processor and careful adjustment of whole circuit. The DPD method has been described in detail in reference, “RF and Microwave Circuit Design for Wireless Communication”, edited by L. E. Larson, Artech House (1996), Chapter 4. 
     The use of the DPD method described above along with the present invention can further improve the efficiency of the output signal from the LA.  FIG. 21  shows a high efficiency RF output power amplifying system incorporating both HGDA-FM-LA combination and DPD in parallel connection. Notice that the input RF signal is an analog signal from the FM and must be converted to a digital signal using the ADC before the RF signal from the FM enters the CFR of the DPD method. The output of the FM is coupled to the CFR to send part of the signal from the FM to the CFR. The signal from the FM to the CFR and DPD is a small percentage of the total signal outputted from the FM, whereby the remaining percentage of the signal is sent to LA through the Adder. The output signal from the LA is coupled to an ADC, such that a small percentage of the total signal outputted from the LA is sent to the ADC, whereby the remaining percentage of the signal is usually sent to an antenna. The signal that travels through the ADC is converted to a digital signal and is inputted to the DPD. The signals from the CFR and ADC are processed by the DPD according to known methods consistent with the DPD method. The end result of the processing by the DPD produces a modified signal that is outputted to a DAC for conversion from a digital signal to an analog signal. A second HGDA is used between the DPD and the LA. The second HGDA is used to amplify the analog signal from the DAC to be the similar strength as the signal from the FM to the Adder. The second HGDA does not necessarily have to be operated near its saturation point in the same manner as the first HGDA. Typically, the gain in signal strength is from 10 to 40 dBs at the second HGDA to achieve proper signal strength to the Adder. The Adder is a known device used to combine two or more signals to form one signal. The signal that is outputted from the Adder produces a modified signal that is sent to the LA. The result is an output signal that has further improved ACLR properties by using the DPD method with the present invention. 
       FIG. 22  depicts a known method Adaptive Feed Forward Linearization (AFL) with the conventional PAs of  FIG. 1  in order to obtain a good quality ACLR output signal. AFL method improves the output signal by using the Inter-Modulation Distortion (IMD) portion of the output RF signal and subtracting an opposite polarity IMD signal that is similar in magnitude. The opposite polarity IMD signal is obtained by processing the signal that enters the PA, prior to that signal entering the PA and feeding the result forward to the output of the PA. The details of the AFL method are described in reference. “RF Microelectronics”, by B. Razavi, Prentice Hall (1998), Chapter 9.  FIG. 22  shows the basics of how the AFL method is employed with a PA. The upward arrows indicate magnitude of the signal. The signal enters the PA have a minimal amount of distortion, as shown by the two upward arrows at  10 . When the signal exits the PA, the signal is increased in magnitude as shown by the two middle arrows at  12 , but the signal also includes distortion as indicated by the shorter arrows on either side of the two middle arrows. The magnitude of the shorter arrows represents the strength of IMD. The signal is delayed by a delay line device for timing. A small percentage of the signal that enters the PA is diverted by a coupler at  14  to a delay line. The two delay lines of the AFL provide proper timing for processing the signal that enters the PA. The signal at  14  is similar to the signal at  10 , but is lower in magnitude. The signal at  14  is sent to an adder. A small percentage of the signal at  12  is sent to an attenuator to reduce the magnitude of the signal taken from the signal at  12 . That signal is sent to the adder. Combining the signals at the adder using subtraction provides a signal at  16  that only includes the distortion portion (IMD) of the signal from  14 . The signal at  16  is inputted to an error amp to increase the magnitude of the IDM signal to produce a signal at  18  which has a similar magnitude to the IMD signal exiting the delay line at  20 . The error amp usually operates linearly with a gain from 10 to 40 dB. The signal at  18  is send to a second adder, as well is the signal at  20 . The signal at  18  is subtracted from the signal at  20  to produce a final output signal at  22  that does not possess the distortion. 
       FIG. 23  shows the use of the AFL method combined with the present invention to further improve the output signal from the LA.  FIG. 23  shows components of the AFL of  FIG. 22  incorporated with the HGDA-FM-LA combination.  FIG. 23  shows a small percentage of the signal from the FM directed to the AFL, along with a small percentage of the signal from the LA to produce an output signal at the second adder that is much improved. There is a connection between the filter module and the first adder to receive and deliver the small percentage of the processed first amplified signal from the filter module to the first adder. The attenuator is connected to the LA to receive a percentage of the second amplified signal from the LA. The attenuator is connected to the first adder to deliver a processed second amplified signal to the first adder. The error amplifier is connected to the first adder to receive a first combined signal which was formed from the processed first amplified signal and processed second amplified signal. The second adder connected to the error amplifier and the LA receive and combine an amplified first combined signal from the error amp and the second amplified signal to produce the output signal. 
     In some communication units, the input signal to be amplified in an amplification system of the communication unit is from a digital source, instead of an analog RF signal from an antenna. For example, the signal to be outputted can be delivered by a fiber optic cable and must eventually be converted to an analog signal for wireless transmission.  FIG. 24  shows the DPD used with HGDA-FM-LA combination. The DPD of  FIG. 24  is the same as the DPD of  FIG. 19 . In the case of  FIG. 24 , the DPD receives a digital input signal as the initial input signal and receives the feedback signal from the HGDA instead of the LA, but in the same manner. The digital input signal in this case does not have to be converted to be used with the DPD and is feed directly to the CFR. Then, the signal is converted to an analog signal and adjusted using up converting frequency mixer that is connected to an LO before reaching the HGDA. The interconnection of the DPD of  FIG. 24  employs the use of a LO and frequency mixing device as shown in  FIG. 19 , instead of the adder shown in  FIG. 21 . The feedback signal is adjusted using down converting frequency mixer before reaching the ADC. In the alternative, the feedback signal can taken from the LA instead of the HGDA. Also, the configuration of  FIG. 24  can be used where an analog RF input signal is converted to a digital form to become the digital input signal and using the HGDA or LA for the taking the feedback signal. 
       FIG. 25  shows the HGDA-FM-LA combination of the present invention combined with the DPD circuit of  FIG. 21  and the AFL circuit of  FIG. 23  to maximize enhancement of the RF power output efficiency of the HGDA-FM-LA combination. Note, both feedback signals for the DPD and AFL are obtained from the output of the LA. The three methods have a similar goal of enhancing the efficiency, but they act on the different locations and the different connections between the input and output of the amplification system. The HGDA-FM is acting on the input side of the LA connected in a series manner. The DPD is acting on the input side of the LA connected in parallel manner, and AFL is acting on output side of LA in a series connection manner. Consequently, all three different techniques having same the goal have a synergy effect enhancing the efficiency of the LA, without the addition of signal interferences among them.  FIG. 26  shows the HGDA-FM-LA combination of the present invention combined with the DPD circuit of  FIG. 24  and the AFL circuit of  FIG. 23 . In  FIG. 26  the DPD is in series and accepts the digital signal, as was described for  FIG. 24 . As was for the embodiment of  FIG. 24 , the feedback signal for the DPD can come from either the HGDA or the LA. 
     The amplification system of the present invention can be combine with a more efficient amplifier, know as the Doherty amplifier. The Doherty amplifier is based on improving the linearity of RF output power amplifier response by combining two complementary amplifiers in parallel manner. Therefore, the Doherty amplifier can be operated under close to an optimum efficiency condition at near its saturation point without significant power spectrum growth of output signal due to the Inter-Modulation Distortion (IMD).  FIG. 27  depicts the schematic design and a graphical representation of how the Doherty amplifier works. The schematic design shows an IN node for an input signal. The signal is split and amplified by a main PA and an auxiliary PA. The signal is then combined for output. The graphical representation shows that the main PA operates near it saturation point, where the power out increases at less of a rate compared to the power in. While, the Auxiliary PA operates such that the power out increases at more a rate as compared to the power in. When signals from the two amplifiers are combined, a signal is produce as shown by the dotted combination line. Detail explanations on this subject can be found in reference, “RF Power Amplifiers for Wireless Communications”, by Steve C. Cripps, Chapter 8, Artech House Inc. 1999. 
       FIG. 28  shows a Doherty amplifier used as the LA, where there the main amplifier and the auxiliary amplifier. The signal is split at the FM and directed to both the main amplifier and the auxiliary amplifier. The outputs from the main amplifier and the auxiliary amplifier are then combined at the adder to produce the output signal to the antenna. Both the main amplifier and the auxiliary amplifier should have the same gain and that gain should be the gain in signal strength desired at the LA position. The Doherty amplifier contributes in two ways when used for the LA. The first way is to enhance the efficiency of the RF power output by improving the linearity of characteristics of an amplifier unit using two complementary amplifiers connected in parallel manner. The second way is to increase level of output power close to twice value of Class B or Class AB power amplifiers with the same gain, because it contains two power amplifiers connected in parallel manner which is one way to increase output power level.  FIG. 29  shows the Doherty amplifier replacing the LA for the DPD and HGDA-FM-LA combination shown in  FIG. 24 .  FIG. 30  shows the Doherty amplifier replacing the LA for the AFL and HGDA-FM-LA combination shown in  FIG. 23 .  FIG. 31  shows the Doherty amplifier replacing the LA for the DPD, AFL and HGDA-FM-LA combination shown in  FIG. 26 . 
     While different embodiment of the invention have been described in detail herein, it will be appreciated by those skilled in the art that various modification and alternatives to embodiments could be developed in light of the overall teachings of the disclosure. Accordingly, the particular arrangements are illustrated only and are not limiting as to the scope of the invention that is to be given the full breadth of any and all equivalents thereof.