Patent Publication Number: US-11646780-B2

Title: Signal generating method and signal generating device

Description:
TECHNICAL FIELD 
     Cross Reference to Related Application 
     This application is based on applications No. 2011-093540 filed in Japan on Apr. 19, 2011 and No. 2011-140749 filed in Japan on Jun. 24, 2011 the contents of which are hereby incorporated by reference. 
     The present invention relates to a signal generating method and a signal generating device for communication using multiple antennas. 
     BACKGROUND ART 
     A MIMO (Multiple-Input, Multiple-Output) system is an example of a conventional communication system using multiple antennas. In multi-antenna communication, of which the MIMO system is typical, multiple transmission signals are each modulated, and each modulated signal is simultaneously transmitted from a different antenna in order to increase the transmission speed of the data. 
       FIG.  23    illustrates a sample configuration of a transmission and reception device having two transmit antennas and two receive antennas, and using two transmit modulated signals (transmit streams). In the transmission device, encoded data are interleaved, the interleaved data are modulated, and frequency conversion and the like are performed to generate transmission signals, which are then transmitted from antennas. In this case, the scheme for simultaneously transmitting different modulated signals from different transmit antennas at the same time and on a common frequency is a spatial multiplexing MIMO system. 
     In this context, Patent Literature 1 suggests using a transmission device provided with a different interleaving pattern for each transmit antenna. That is, the transmission device from  FIG.  23    should use two distinct interleaving patterns performed by two interleavers (πa and πb). As for the reception device, Non-Patent Literature 1 and Non-Patent Literature 2 describe improving reception quality by iteratively using soft values for the detection method (by the MIMO detector of  FIG.  23   ). 
     As it happens, models of actual propagation environments in wireless communications include NLOS (Non Line-Of-Sight), typified by a Rayleigh fading environment, and LOS (Line-Of-Sight), typified by a Rician fading environment. When the transmission device transmits a single modulated signal, and the reception device performs maximal ratio combination on the signals received by a plurality of antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment, in particular in an environment where the Rician factor is large. The Rician factor represents the received power of direct waves relative to the received power of scattered waves. However, depending on the transmission system (e.g., a spatial multiplexing MIMO system), a problem occurs in that the reception quality deteriorates as the Rician factor increases (see Non-Patent Literature 3). 
       FIGS.  24 A and  24 B  illustrate an example of simulation results of the BER (Bit Error Rate) characteristics (vertical axis: BER, horizontal axis: SNR (signal-to-noise ratio) for data encoded with LDPC (low-density parity-check) codes and transmitted over a 2×2 (two transmit antennas, two receive antennas) spatial multiplexing MIMO system in a Rayleigh fading environment and in a Rician fading environment with Rician factors of K=3, 10, and 16 dB.  FIG.  24 A  gives the Max-Log approximation-based log-likelihood ratio (i.e., Max-log APP, where APP is the a posteriori probability) BER characteristics without iterative detection (see Non-Patent Literature 1 and Non-Patent Literature 2), while  FIG.  24 B  gives the Max-log APP BER characteristic with iterative detection (see Non-Patent Literature 1 and Non-Patent Literature 2) (number of iterations: five).  FIGS.  24 A and  24 B  clearly indicate that, regardless of whether or not iterative detection is performed, reception quality degrades in the spatial multiplexing MIMO system as the Rician factor increases. Thus, the problem of reception quality degradation upon stabilization of the propagation environment in the spatial multiplexing MIMO system, which does not occur in a conventional single-modulation signal system, is unique to the spatial multiplexing MIMO system. 
     Broadcast or multicast communication is a service applied to various propagation environments. The radio wave propagation environment between the broadcaster and the receivers belonging to the users is often a LOS environment. When using a spatial multiplexing MIMO system having the above problem for broadcast or multicast communication, a situation may occur in which the received electric field strength is high at the reception device, but in which degradation in reception quality makes service reception impossible. In other words, in order to use a spatial multiplexing MIMO system in broadcast or multicast communication in both the NLOS environment and the LOS environment, a MIMO system that offers a certain degree of reception quality is desirable. 
     Non-Patent Literature 8 describes a method of selecting a codebook used in precoding (i.e. a precoding matrix, also referred to as a precoding weight matrix) based on feedback information from a communication party. However, Non-Patent Literature 8 does not at all disclose a method for precoding in an environment in which feedback information cannot be acquired from the other party, such as in the above broadcast or multicast communication. 
     On the other hand, Non-Patent Literature 4 discloses a method for switching the precoding matrix over time. This method is applicable when no feedback information is available. Non-Patent Literature 4 discloses using a unitary matrix as the precoding matrix, and switching the unitary matrix at random, but does not at all disclose a method applicable to degradation of reception quality in the above-described LOS environment. Non-Patent Literature 4 simply recites hopping between precoding matrices at random. Obviously, Non-Patent Literature 4 makes no mention whatsoever of a precoding method, or a structure of a precoding matrix, for remedying degradation of reception quality in a LOS environment. 
     CITATION LIST 
     Patent Literature 
     [Patent Literature 1] 
     
         
         International Patent Application Publication No. WO2005/050885 
       
    
     Non-Patent Literature 
     [Non-Patent Literature 1] 
     
         
         “Achieving near-capacity on a multiple-antenna channel” IEEE Transaction on communications, vol. 51, no. 3, pp. 389-399, March 2003
 
[Non-Patent Literature 2] “Performance analysis and design optimization of LDPC-coded MIMO OFDM systems” IEEE Trans. Signal Processing, vol. 52, no. 2, pp. 348-361, February 2004
 
[Non-Patent Literature 3]
 
         “BER performance evaluation in 2×2 MIMO spatial multiplexing systems under Rician fading channels” IEICE Trans. Fundamentals, vol. E91-A, no. 10, pp. 2798-2807, October 2008
 
[Non-Patent Literature 4]
 
         “Turbo space-time codes with time varying linear transformations” IEEE Trans. Wireless communications, vol. 6, no. 2, pp. 486-493, February 2007
 
[Non-Patent Literature 5]
 
         “Likelihood function for QR-MLD suitable for soft-decision turbo decoding and its performance” IEICE Trans. Commun., vol. E88-B, no. 1, pp. 47-57, January 2004
 
[Non-Patent Literature 6]
 
         “A tutorial on ‘Parallel concatenated (Turbo) coding’, ‘Turbo (iterative) decoding’ and related topics” IEICE, Technical Report IT98-51
 
[Non-Patent Literature 7]
 
         “Advanced signal processing for PLCs: Wavelet-OFDM” Proc. of IEEE International symposium on ISPLC 2008, pp. 187-192, 2008
 
[Non-Patent Literature 8]
 
         D. J. Love and R. W. Heath Jr., “Limited feedback unitary precoding for spatial multiplexing systems” IEEE Trans. Inf. Theory, vol. 51, no. 8, pp. 2967-2976, August 2005
 
[Non-Patent Literature 9]
 
         DVB Document A122, Framing structure, channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), June 2008
 
[Non-Patent Literature 10]
 
         L. Vangelista, N. Benvenuto, and S. Tomasin “Key technologies for next-generation terrestrial digital television standard DVB-T2,” IEEE Commun. Magazine, vol. 47, no. 10, pp. 146-153, October 2009
 
[Non-Patent Literature 11]
 
         T. Ohgane, T. Nishimura, and Y. Ogawa, “Application of space division multiplexing and those performance in a MIMO channel” IEICE Trans. Commun., vol. E88-B, no. 5, pp. 1843-1851, May 2005
 
[Non-Patent Literature 12]
 
         R. G. Gallager “Low-density parity-check codes,” IRE Trans. Inform. Theory, IT-8, pp. 21-28, 1962
 
[Non-Patent Literature 13]
 
         D. J. C. Mackay, “Good error-correcting codes based on very sparse matrices,” IEEE Trans. Inform. Theory, vol. 45, no. 2, pp. 399-431, March 1999.
 
[Non-Patent Literature 14]
 
         ETSI EN 302 307, “Second generation framing structure, channel coding and modulation systems for broadcasting, interactive services, news gathering and other broadband satellite applications” v.1.1.2, June 2006
 
[Non-Patent Literature 15]
 
         Y.-L. Ueng, and C.-C. Cheng “A fast-convergence decoding method and memory-efficient VLSI decoder architecture for irregular LDPC codes in the IEEE 802.16e standards” IEEE VTC-2007 Fall, pp. 1255-1259
 
[Non-Patent Literature 16]
 
         S. M. Alamouti “A simple transmit diversity technique for wireless communications” IEEE J. Select. Areas Commun., vol. 16, no. 8, pp. 1451-1458, October 1998
 
[Non-Patent Literature 17]
 
         V. Tarokh, H. Jafrkhani, and A. R. Calderbank “Space-time block coding for wireless communications: Performance results” IEEE J. Select. Areas Commun., vol. 17, no. 3, no. 3, pp. 451-460, March 1999 
       
    
     SUMMARY OF INVENTION 
     Technical Problem 
     An object of the present invention is to provide a MIMO system that improves reception quality in a LOS environment. 
     Solution to Problem 
     The present invention provides a signal generation method for generating, from a plurality of baseband signals, a plurality of signals for transmission on a common frequency band and at a common time, comprising: performing a change of phase on each of a first baseband signal s 1  generated from a first set of bits according to a first modulation scheme and a second baseband signal s 2  generated from a second set of bits according to a second modulation scheme, thus generating a first post-phase-change baseband signal s 1 ′ and a second post-phase-change baseband signal s 2 ′; multiplying the first post-phase-change baseband signal s 1 ′ by u and multiplying the second post-phase-change baseband signal s 2 ′ by v, where u and v denote real numbers different from each other; and applying weighting according to a predetermined matrix F to the first post-phase-change baseband signal s 1 ′×u and to the second post-phase-change baseband signal s 2 ′×v, thus generating the plurality of signals for transmission on the common frequency band and at the common time as a first weighted signal z 1  and a second weighted signal z 2 , wherein the first weighted signal z 1  and the second weighted signal z 2  satisfy the relation: (z 1 , z 2 ) T =F(u×s 1 ′, v×s 2 ′) T , and the first modulation scheme is different from the second modulation scheme. 
     The present invention also provides a signal generation apparatus for generating, from a plurality of baseband signals, a plurality of signals for transmission on a common frequency band and at a common time, comprising: a phase changer performing a change of phase on each of a first baseband signal s 1  generated from a first set of bits according to a first modulation scheme and a second baseband signal s 2  generated from a second set of bits according to a second modulation scheme, thus generating a first post-phase-change baseband signal s 1 ′ and a second post-phase-change baseband signal s 2 ′; a power changer multiplying the first post-phase-change baseband signal s 1 ′ by u and multiplying the second post-phase-change baseband signal s 2 ′ by v, where u and v denote real numbers different from each other; and a weighting unit applying weighting according to a predetermined matrix F to the first post-phase-change baseband signal s 1 ′×u and to the second post-phase-change baseband signal s 2 ′×v, thus generating the plurality of signals for transmission on the common frequency band and at the common time as a first weighted signal z 1  and a second weighted signal z 2 , wherein the first weighted signal z 1  and the second weighted signal z 2  satisfy the relation: (z 1 , z 2 ) T =F(u×s 1 ′,v×s 2 ′) T , and the first modulation scheme is different from the second modulation scheme. 
     Advantageous Effects of Invention 
     According to the above structure, the present invention provides a signal generation method and a signal generation apparatus that remedy degradation of reception quality in a LOS environment, thereby providing high-quality service to LOS users during broadcast or multicast communication. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG.  1    illustrates an example of a transmission and reception device in a spatial multiplexing MIMO system. 
         FIG.  2    illustrates a sample frame configuration. 
         FIG.  3    illustrates an example of a transmission device applying a phase changing method. 
         FIG.  4    illustrates another example of a transmission device applying a phase changing method. 
         FIG.  5    illustrates another sample frame configuration. 
         FIG.  6    illustrates another sample phase changing method. 
         FIG.  7    illustrates a sample configuration of a reception device. 
         FIG.  8    illustrates a sample configuration of a signal processor in the reception device. 
         FIG.  9    illustrates another sample configuration of a signal processor in the reception device. 
         FIG.  10    illustrates an iterative decoding method. 
         FIG.  11    illustrates sample reception conditions. 
         FIG.  12    illustrates a further example of a transmission device applying a phase changing method. 
         FIG.  13    illustrates yet a further example of a transmission device applying a phase changing method. 
         FIGS.  14 A and  14 B  illustrate another sample frame configuration. 
         FIGS.  15 A and  15 B  illustrate another sample frame configuration. 
         FIGS.  16 A and  16 B  illustrate another sample frame configuration. 
         FIGS.  17 A and  17 B  illustrate another sample frame configuration. 
         FIGS.  18 A and  18 B  illustrate another sample frame configuration. 
         FIGS.  19 A and  19 B  illustrate examples of a mapping method. 
         FIGS.  20 A and  20 B  illustrate further examples of a mapping method. 
         FIG.  21    illustrates a sample configuration of a weighting unit. 
         FIG.  22    illustrates a sample symbol rearrangement method. 
         FIG.  23    illustrates another example of a transmission and reception device in a spatial multiplexing MIMO system. 
         FIGS.  24 A and  24 B  illustrate sample BER characteristics. 
         FIG.  25    illustrates another sample phase changing method. 
         FIG.  26    illustrates another sample phase changing method. 
         FIG.  27    illustrates another sample phase changing method. 
         FIG.  28    illustrates another sample phase changing method. 
         FIG.  29    illustrates another sample phase changing method. 
         FIG.  30    illustrates a sample symbol arrangement for a modulated signal providing high received signal quality. 
         FIG.  31    illustrates a sample frame configuration for a modulated signal providing high received signal quality. 
         FIG.  32    illustrates a sample symbol arrangement for a modulated signal providing high received signal quality. 
         FIG.  33    illustrates a sample symbol arrangement for a modulated signal providing high received signal quality. 
         FIG.  34    illustrates a variation in numbers of symbols and slots needed per pair of coded blocks when block codes are used. 
         FIG.  35    illustrates another variation in numbers of symbols and slots needed per pair of coded blocks when block codes are used. 
         FIG.  36    illustrates an overall configuration of a digital broadcasting system. 
         FIG.  37    is a block diagram illustrating a sample receiver. 
         FIG.  38    illustrates multiplexed data configuration. 
         FIG.  39    is a schematic diagram illustrating multiplexing of encoded data into streams. 
         FIG.  40    is a detailed diagram illustrating a video stream as contained in a PES packet sequence. 
         FIG.  41    is a structural diagram of TS packets and source packets in the multiplexed data. 
         FIG.  42    illustrates PMT data configuration. 
         FIG.  43    illustrates information as configured in the multiplexed data. 
         FIG.  44    illustrates the configuration of stream attribute information. 
         FIG.  45    illustrates the configuration of a video display and audio output device. 
         FIG.  46    illustrates a sample configuration of a communications system. 
         FIGS.  47 A and  47 B  illustrate sample symbol arrangements for a modulated signal providing high received signal quality. 
         FIGS.  48 A and  48 B  illustrate sample symbol arrangements for a modulated signal providing high received signal quality. 
         FIGS.  49 A and  49 B  illustrate sample symbol arrangements for a modulated signal providing high received signal quality. 
         FIGS.  50 A and  50 B  illustrate sample symbol arrangements for a modulated signal providing high received signal quality. 
         FIG.  51    illustrates a sample configuration of a transmission device. 
         FIG.  52    illustrates another sample configuration of a transmission device. 
         FIG.  53    illustrates a further sample configuration of a transmission device. 
         FIG.  54    illustrates yet a further sample configuration of a transmission device. 
         FIG.  55    illustrates a baseband signal switcher. 
         FIG.  56    illustrates yet still a further sample configuration of a transmission device. 
         FIG.  57    illustrates sample operations of a distributor. 
         FIG.  58    illustrates further sample operations of a distributor. 
         FIG.  59    illustrates a sample communications system indicating the relationship between base stations and terminals. 
         FIG.  60    illustrates an example of transmit signal frequency allocation. 
         FIG.  61    illustrates another example of transmit signal frequency allocation. 
         FIG.  62    illustrates a sample communications system indicating the relationship between a base station, repeaters, and terminals. 
         FIG.  63    illustrates an example of transmit signal frequency allocation with respect to the base station. 
         FIG.  64    illustrates an example of transmit signal frequency allocation with respect to the repeaters. 
         FIG.  65    illustrates a sample configuration of a receiver and transmitter in the repeater. 
         FIG.  66    illustrates a signal data format used for transmission by the base station. 
         FIG.  67    illustrates yet still another sample configuration of a transmission device. 
         FIG.  68    illustrates another baseband signal switcher. 
         FIG.  69    illustrates a sample weighting, baseband signal switching, and phase changing method. 
         FIG.  70    illustrates a sample configuration of a transmission device using an OFDM method. 
         FIGS.  71 A and  71 B  illustrate another sample frame configuration. 
         FIG.  72    further illustrates the numbers of slots and phase changing values corresponding to a modulation scheme. 
         FIG.  73    further illustrates the numbers of slots and phase changing values corresponding to a modulation scheme. 
         FIG.  74    illustrates the overall frame configuration of a signal transmitted by a broadcaster using DVB-T2. 
         FIG.  75    illustrates two or more types of signals at the same timestamp. 
         FIG.  76    illustrates still a further sample configuration of a transmission device. 
         FIG.  77    illustrates an alternate sample frame configuration. 
         FIG.  78    illustrates another alternate sample frame configuration. 
         FIG.  79    illustrates a further alternate sample frame configuration. 
         FIG.  80    illustrates yet a further alternate sample frame configuration. 
         FIG.  81    illustrates yet another alternate sample frame configuration. 
         FIG.  82    illustrates still another alternate sample frame configuration. 
         FIG.  83    illustrates still a further alternate sample frame configuration. 
         FIG.  84    further illustrates two or more types of signals at the same timestamp. 
         FIG.  85    illustrates an alternate sample configuration of a transmission device. 
         FIG.  86    illustrates an alternate sample configuration of a reception device. 
         FIG.  87    illustrates another alternate sample configuration of a reception device. 
         FIG.  88    illustrates yet another alternate sample configuration of a reception device. 
         FIGS.  89 A and  89 B  illustrate further alternate sample frame configurations. 
         FIGS.  90 A and  90 B  illustrate yet further alternate sample frame configurations. 
         FIGS.  91 A and  91 B  illustrate more alternate sample frame configurations. 
         FIGS.  92 A and  92 B  illustrate yet more alternate sample frame configurations. 
         FIGS.  93 A and  93 B  illustrate still further alternate sample frame configurations. 
         FIG.  94    illustrates a sample frame configuration used when space-time block codes are employed. 
         FIG.  95    illustrates an example of signal point distribution for 16-QAM in the I-Q plane. 
         FIG.  96    illustrates an example of signal point distribution for QPSK in the I-Q plane. 
         FIG.  97    schematically shows absolute values of a log-likelihood ratio obtained by the reception device. 
         FIG.  98    schematically shows absolute values of a log-likelihood ratio obtained by the reception device. 
         FIG.  99    is an example of a structure of a signal processor pertaining to a weighting unit. 
         FIG.  100    is an example of a structure of the signal processor pertaining to the weighting unit. 
         FIG.  101    illustrates an example of signal point distribution for 64-QAM in the I-Q plane. 
         FIG.  102    illustrates an example of signal point distribution for 16-QAM in the I-Q plane. 
         FIG.  103    indicates a sample configuration for a signal generator when cyclic Q delay is applied. 
         FIG.  104    illustrates a first example of a generation method for s 1 ( t ) and s 2 ( t ) when cyclic Q delay is used. 
         FIG.  105    indicates a sample configuration for a signal generator when cyclic Q delay is applied. 
         FIG.  106    indicates a sample configuration for a signal generator when cyclic Q delay is applied. 
         FIG.  107    illustrates a second example of a generation method for s 1 ( t ) and s 2 ( t ) when cyclic Q delay is used. 
         FIG.  108    indicates a sample configuration for a signal generator when cyclic Q delay is applied. 
         FIG.  109    indicates a sample configuration for a signal generator when cyclic Q delay is applied. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Embodiments of the present invention are described below with reference to the accompanying drawings. 
     Embodiment 1 
     The following describes, in detail, a transmission method, a transmission device, a reception method, and a reception device pertaining to the present Embodiment. 
     Before beginning the description proper, an outline of transmission schemes and decoding schemes in a conventional spatial multiplexing MIMO system is provided. 
       FIG.  1    illustrates the structure of an Nt×Nr spatial multiplexing MIMO system. An information vector z is encoded and interleaved. The encoded bit vector u=(u 1 , . . . , u Nt ) is obtained as the interleave output. Here, u i =(u i1 , . . . , u iM ) (where M is the number of transmitted bits per symbol). For a transmit vector s=(s 1 , . . . , S Nt ), a received signal s i =map(u i ) is found for transmit antenna #i. Normalizing the transmit energy, this is expressible as E{|s i | 2 }=E s /Nt (where E s  is the total energy per channel). The receive vector y=(y 1 , . . . , y Nr ) T  is expressed in Math. 1 (formula 1), below. 
     
       
         
           
             
               
                 
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                   [     Math   .         3     ]                                     L   ⁡   (   u   )     =       (       L   ⁡   (     u   1     )     ,   …          ,     L   ⁡   (     u     N   t       )       )     T             (     formula   ⁢         3     )                           [     Math   .         4     ]                                     L   ⁡   (     u   i     )     =     (       L   ⁡   (     u     i   ⁢   1       )     ,   …          ,     L   ⁡   (     u   iM     )       )             (     formula   ⁢         4     )                           [     Math   .        5     ]                                     L   ⁡   (     u   ij     )     =     ln   ⁢       P   ⁡   (       u   ij     =     +   1       )       P   ⁡   (       u   ij     =     -   1       )                 (     formula   ⁢         5     )               
(Iterative Detection Method)
 
     The following describes the MIMO signal iterative detection performed by the N t ×N r  spatial multiplexing MIMO system. The log-likelihood ratio of u mn  is defined by Math. 6 (formula 6). 
     
       
         
           
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                                         ❘ 
                                         
                                           u 
                                           mn 
                                         
                                       
                                       = 
                                       
                                         + 
                                         1 
                                       
                                     
                                     ) 
                                   
                                   
                                     p 
                                     ⁡ 
                                     ( 
                                     
                                       
                                         y 
                                         ❘ 
                                         
                                           u 
                                           mn 
                                         
                                       
                                       = 
                                       
                                         - 
                                         1 
                                       
                                     
                                     ) 
                                   
                                 
                               
                             
                           
                         
                       
                       
                         
                           
                             = 
                             
                               
                                 ln 
                                 ⁢ 
                                 
                                   
                                     P 
                                     ⁡ 
                                     ( 
                                     
                                       
                                         u 
                                         mn 
                                       
                                       = 
                                       
                                         + 
                                         1 
                                       
                                     
                                     ) 
                                   
                                   
                                     P 
                                     ⁡ 
                                     ( 
                                     
                                       
                                         u 
                                         mn 
                                       
                                       = 
                                       
                                         - 
                                         1 
                                       
                                     
                                     ) 
                                   
                                 
                               
                               + 
                               
                                 ln 
                                 ⁢ 
                                 
                                   
                                     
                                       Σ 
                                       
                                         U 
                                         
                                           mn 
                                           , 
                                           
                                             + 
                                             1 
                                           
                                         
                                       
                                     
                                     ⁢ 
                                     
                                       p 
                                       ⁡ 
                                       ( 
                                       
                                         y 
                                         ❘ 
                                         u 
                                       
                                       ) 
                                     
                                     ⁢ 
                                     
                                       p 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ❘ 
                                         
                                           u 
                                           mn 
                                         
                                       
                                       ) 
                                     
                                   
                                   
                                     
                                       Σ 
                                       
                                         U 
                                         
                                           mn 
                                           , 
                                           
                                             - 
                                             1 
                                           
                                         
                                       
                                     
                                     ⁢ 
                                     
                                       p 
                                       ⁡ 
                                       ( 
                                       
                                         y 
                                         ❘ 
                                         u 
                                       
                                       ) 
                                     
                                     ⁢ 
                                     
                                       p 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ❘ 
                                         
                                           u 
                                           mn 
                                         
                                       
                                       ) 
                                     
                                   
                                 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       ( 
                       
                         formula 
                         ⁢ 
                             
                         7 
                       
                       ) 
                     
                   
                 
               
             
           
         
       
     
     Note that U mn, ±1 ={u|u mn =±1}. Through the approximation ln Σaj˜max ln a j , Math. 7 (formula 7) can be approximated as Math. 8 (formula 8). The symbol ˜ is herein used to signify approximation. 
     
       
         
           
                             
             
               [ 
               
                 Math 
                 . 
                     
                 8 
               
               ] 
             
           
         
       
       
         
           
             
               
                 
                   
 
                   
                     
                       L 
                       ⁡ 
                       ( 
                       
                         
                           u 
                           mn 
                         
                         ❘ 
                         y 
                       
                       ) 
                     
                     ≈ 
                     
                       
                         ln 
                         ⁢ 
                         
                           
                             P 
                             ⁡ 
                             ( 
                             
                               
                                 u 
                                 mn 
                               
                               = 
                               
                                 + 
                                 1 
                               
                             
                             ) 
                           
                           
                             P 
                             ⁡ 
                             ( 
                             
                               
                                 u 
                                 mn 
                               
                               = 
                               
                                 - 
                                 1 
                               
                             
                             ) 
                           
                         
                       
                       + 
                     
                   
                 
               
               
                 
                   
                     ( 
                     
                       formula 
                       ⁢ 
                           
                       8 
                     
                     ) 
                   
                   TagBox[RowBox[List[&#34;(&#34;, RowBox[List[&#34;formula&#34;, &#34;   &#34;, &#34;8&#34;]], &#34;)&#34;]], Null, Rule[Editable, True], Rule[Selectable, True]] 
                 
               
             
           
         
       
       
         
           
                                        
             
               
                 
                   max 
                   
                     Umn 
                     , 
                     
                       + 
                       1 
                     
                   
                 
                 
                   { 
                   
                     
                       ln 
                       ⁢ 
                           
                       
                         p 
                         ⁡ 
                         ( 
                         
                           y 
                           ❘ 
                           u 
                         
                         ) 
                       
                     
                     + 
                     
                       P 
                       ⁡ 
                       ( 
                       
                         u 
                         ❘ 
                         
                           u 
                           mn 
                         
                       
                       ) 
                     
                   
                   } 
                 
               
               - 
               
                 
                   max 
                   
                     Umn 
                     , 
                     
                       - 
                       1 
                     
                   
                 
                 
                   { 
                   
                     
                       ln 
                       ⁢ 
                           
                       
                         p 
                         ⁡ 
                         ( 
                         
                           y 
                           ❘ 
                           u 
                         
                         ) 
                       
                     
                     + 
                     
                       P 
                       ⁡ 
                       ( 
                       
                         u 
                         ❘ 
                         
                           u 
                           mn 
                         
                       
                       ) 
                     
                   
                   } 
                 
               
             
           
         
       
     
     In Math. 8 (formula 8), P(u|u mn ) and ln P(u|u mn ) can be expressed as follows. 
     
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       9 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       
                         
                           
                             P 
                             ⁡ 
                             ( 
                             
                               u 
                               ❘ 
                               
                                 u 
                                 mn 
                               
                             
                             ) 
                           
                           = 
                           
                             
                               Π 
                               
                                 
                                   ( 
                                   ij 
                                   ) 
                                 
                                 ≠ 
                                 
                                   ( 
                                   mn 
                                   ) 
                                 
                               
                             
                             ⁢ 
                             
                               P 
                               ⁡ 
                               ( 
                               
                                 u 
                                 ij 
                               
                               ) 
                             
                           
                         
                       
                     
                     
                       
                         
                           = 
                           
                             
                               Π 
                               
                                 
                                   ( 
                                   ij 
                                   ) 
                                 
                                 ≠ 
                                 
                                   ( 
                                   mn 
                                   ) 
                                 
                               
                             
                             ⁢ 
                             
                               
                                 exp 
                                 ⁡ 
                                 ( 
                                 
                                   
                                     
                                       u 
                                       ij 
                                     
                                     ⁢ 
                                     
                                       L 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                   
                                   2 
                                 
                                 ) 
                               
                               
                                 
                                   exp 
                                   ⁡ 
                                   ( 
                                   
                                     
                                       L 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                     2 
                                   
                                   ) 
                                 
                                 + 
                                 
                                   exp 
                                   ⁡ 
                                   ( 
                                   
                                     - 
                                     
                                       
                                         L 
                                         ⁡ 
                                         ( 
                                         
                                           u 
                                           ij 
                                         
                                         ) 
                                       
                                       2 
                                     
                                   
                                   ) 
                                 
                               
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     9 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       10 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       ln 
                       ⁢ 
                       
                         P 
                         ⁡ 
                         ( 
                         
                           u 
                           ❘ 
                           
                             u 
                             mn 
                           
                         
                         ) 
                       
                     
                     = 
                     
                       
                         ( 
                         
                           
                             ∑ 
                             ij 
                           
                           
                             ln 
                             ⁢ 
                             
                               P 
                               ⁡ 
                               ( 
                               
                                 u 
                                 ij 
                               
                               ) 
                             
                           
                         
                         ) 
                       
                       - 
                       
                         ln 
                         ⁢ 
                         
                           P 
                           ⁡ 
                           ( 
                           
                             u 
                             mn 
                           
                           ) 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     10 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       11 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                   
                     
                       
                         
                           ln 
                           ⁢ 
                               
                           
                             P 
                             ⁡ 
                             ( 
                             
                               u 
                               ij 
                             
                             ) 
                           
                         
                           
                         = 
                         
                           
                             
                               1 
                               2 
                             
                             ⁢ 
                             
                               u 
                               ij 
                             
                             ⁢ 
                             
                               P 
                               ⁡ 
                               ( 
                               
                                 u 
                                 ij 
                               
                               ) 
                             
                           
                           - 
                           
                             ln 
                             ⁡ 
                             ( 
                             
                               
                                 exp 
                                 ⁡ 
                                 ( 
                                 
                                   
                                     L 
                                     ⁡ 
                                     ( 
                                     
                                       u 
                                       ij 
                                     
                                     ) 
                                   
                                   2 
                                 
                                 ) 
                               
                               + 
                               
                                 exp 
                                 ⁡ 
                                 ( 
                                 
                                   - 
                                   
                                     
                                       L 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                     2 
                                   
                                 
                                 ) 
                               
                             
                             ) 
                           
                         
                       
                     
                   
                   
                     
                       
                           
                         
                           ≈ 
                           
                             
                               
                                 1 
                                 2 
                               
                               ⁢ 
                               
                                 u 
                                 ij 
                               
                               ⁢ 
                               
                                 L 
                                 ⁡ 
                                 ( 
                                 
                                   u 
                                   ij 
                                 
                                 ) 
                               
                             
                             - 
                             
                               
                                 1 
                                 2 
                               
                               ⁢ 
                               
                                 
                                   ❘ 
                                   &#34;\[LeftBracketingBar]&#34; 
                                 
                                 
                                   L 
                                   ⁡ 
                                   ( 
                                   
                                     u 
                                     ij 
                                   
                                   ) 
                                 
                                 
                                   ❘ 
                                   &#34;\[RightBracketingBar]&#34; 
                                 
                               
                               ⁢ 
                                   
                               for 
                               ⁢ 
                                   
                               
                                 
                                   ❘ 
                                   &#34;\[LeftBracketingBar]&#34; 
                                 
                                 
                                   L 
                                   ⁡ 
                                   ( 
                                   
                                     u 
                                     ij 
                                   
                                   ) 
                                 
                                 
                                   ❘ 
                                   &#34;\[RightBracketingBar]&#34; 
                                 
                               
                             
                           
                           &gt; 
                           2 
                         
                       
                     
                   
                   
                     
                       
                           
                         
                           = 
                           
                             
                               
                                 ❘ 
                                 &#34;\[LeftBracketingBar]&#34; 
                               
                               
                                 
                                   L 
                                   ⁡ 
                                   ( 
                                   
                                     u 
                                     ij 
                                   
                                   ) 
                                 
                                 2 
                               
                               
                                 ❘ 
                                 &#34;\[RightBracketingBar]&#34; 
                               
                             
                             ⁢ 
                             
                               ( 
                               
                                 
                                   
                                     u 
                                     ij 
                                   
                                   ⁢ 
                                   
                                     sign 
                                     ( 
                                     
                                       L 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                     ) 
                                   
                                 
                                 - 
                                 1 
                               
                               ) 
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     11 
                   
                   ) 
                 
               
             
           
         
       
     
     Note that the log-probability of the equation given in Math. 2 (formula 2) can be expressed as Math. 12 (formula 12). 
     
       
         
           
             [ 
             
               Math 
               . 
                   
               12 
             
             ] 
           
         
       
       
         
           
             
               
                 
                   
                     ln 
                     ⁢ 
                     
                       P 
                       ⁡ 
                       ( 
                       
                         y 
                         ❘ 
                         u 
                       
                       ) 
                     
                   
                   = 
                   
                     
                       
                         - 
                         
                           
                             N 
                             r 
                           
                           2 
                         
                       
                       ⁢ 
                       
                         ln 
                         ⁡ 
                         ( 
                         
                           2 
                           ⁢ 
                           
                             πσ 
                             2 
                           
                         
                         ) 
                       
                     
                     - 
                     
                       
                         1 
                         
                           2 
                           ⁢ 
                           
                             σ 
                             2 
                           
                         
                       
                       ⁢ 
                       
                         
                            
                           
                             y 
                             - 
                             
                               Hs 
                               ⁡ 
                               ( 
                               u 
                               ) 
                             
                           
                            
                         
                         2 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     12 
                   
                   ) 
                 
               
             
           
         
       
     
     Accordingly, given Math. 7 (formula 7) and Math. 13 (formula 13), the posterior L-value for the MAP or APP (a posteriori probability) can be can be expressed as follows. 
     
       
         
           
                             
             
               
                 [ 
                 
                   Math 
                   . 
                       
                   13 
                 
                 ] 
               
               ⁢ 
               
 
               
                 
                   
                     
                       
                         L 
                         ⁡ 
                         ( 
                         
                           
                             u 
                             mn 
                           
                           ❘ 
                           y 
                         
                         ) 
                       
                       = 
                       
                         ln 
                         ⁢ 
                         
                           
                             
                               Σ 
                               
                                 U 
                                 
                                   mn 
                                   , 
                                   
                                     + 
                                     1 
                                   
                                 
                               
                             
                             ⁢ 
                             exp 
                             ⁢ 
                             
                               { 
                               
                                 
                                   
                                     - 
                                     
                                       1 
                                       
                                         2 
                                         ⁢ 
                                         
                                           σ 
                                           2 
                                         
                                       
                                     
                                   
                                   ⁢ 
                                   
                                     
                                        
                                       
                                         y 
                                         - 
                                         
                                           Hs 
                                           ⁡ 
                                           ( 
                                           u 
                                           ) 
                                         
                                       
                                        
                                     
                                     2 
                                   
                                 
                                 + 
                                 
                                   
                                     ∑ 
                                     ij 
                                   
                                   
                                     ln 
                                     ⁢ 
                                     
                                       P 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                   
                                 
                               
                               } 
                             
                           
                           
                             
                               Σ 
                               
                                 U 
                                 
                                   mn 
                                   , 
                                   
                                     - 
                                     1 
                                   
                                 
                               
                             
                             ⁢ 
                             exp 
                             ⁢ 
                             
                               { 
                               
                                 
                                   
                                     - 
                                     
                                       1 
                                       
                                         2 
                                         ⁢ 
                                         
                                           σ 
                                           2 
                                         
                                       
                                     
                                   
                                   ⁢ 
                                   
                                     
                                        
                                       
                                         y 
                                         - 
                                         
                                           Hs 
                                           ⁡ 
                                           ( 
                                           u 
                                           ) 
                                         
                                       
                                        
                                     
                                     2 
                                   
                                 
                                 + 
                                 
                                   
                                     ∑ 
                                     ij 
                                   
                                   
                                     ln 
                                     ⁢ 
                                     
                                       P 
                                       ⁡ 
                                       ( 
                                       
                                         u 
                                         ij 
                                       
                                       ) 
                                     
                                   
                                 
                               
                               } 
                             
                           
                         
                       
                     
                   
                   
                     
                       ( 
                       
                         formula 
                         ⁢ 
                             
                         13 
                       
                       ) 
                     
                   
                 
               
             
           
         
       
     
     This is hereinafter termed iterative APP decoding. Also, given Math. 8 (formula 8) and Math. 12 (formula 12), the posterior L-value for the Max-log APP can be can be expressed as follows. 
     
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       14 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       L 
                       ⁡ 
                       ( 
                       
                         
                           u 
                           mn 
                         
                         ❘ 
                         y 
                       
                       ) 
                     
                     ≈ 
                     
                       
                         
                           max 
                           
                             Umn 
                             , 
                             
                               + 
                               1 
                             
                           
                         
                         
                           { 
                           
                             Ψ 
                             ⁡ 
                             ( 
                             
                               u 
                               , 
                               y 
                               , 
                               
                                 L 
                                 ⁡ 
                                 ( 
                                 u 
                                 ) 
                               
                             
                             ) 
                           
                           } 
                         
                       
                       - 
                       
                         
                           max 
                           
                             Umn 
                             , 
                             
                               - 
                               1 
                             
                           
                         
                         
                           { 
                           
                             Ψ 
                             ⁡ 
                             ( 
                             
                               u 
                               , 
                               y 
                               , 
                               
                                 L 
                                 ⁡ 
                                 ( 
                                 u 
                                 ) 
                               
                             
                             ) 
                           
                           } 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     14 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       15 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       Ψ 
                       ⁡ 
                       ( 
                       
                         u 
                         , 
                         y 
                         , 
                         
                           L 
                           ⁡ 
                           ( 
                           u 
                           ) 
                         
                       
                       ) 
                     
                     = 
                     
                       
                         
                           - 
                           
                             1 
                             
                               2 
                               ⁢ 
                               
                                 σ 
                                 2 
                               
                             
                           
                         
                         ⁢ 
                         
                           
                              
                             
                               y 
                               - 
                               
                                 Hs 
                                 ⁡ 
                                 ( 
                                 u 
                                 ) 
                               
                             
                              
                           
                           2 
                         
                       
                       + 
                       
                         
                           ∑ 
                           ij 
                         
                         
                           ln 
                           ⁢ 
                           
                             P 
                             ⁡ 
                             ( 
                             
                               u 
                               ij 
                             
                             ) 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     15 
                   
                   ) 
                 
               
             
           
         
       
     
     This is hereinafter referred to as iterative Max-log APP decoding. As such, the external information required by the iterative decoding system is obtainable by subtracting prior input from Math. 13 (formula 13) or from Math. 14 (formula 14). 
     (System Model) 
       FIG.  23    illustrates the basic configuration of a system related to the following explanations. The illustrated system is a 2×2 spatial multiplexing MIMO system having an outer decoder for each of two streams A and B. The two outer decoders perform identical LDPC encoding. (Although the present example considers a configuration in which the outer encoders use LDPC codes, the outer encoders are not restricted to the use of LDPC as the error-correcting codes. The example may also be realized using other error-correcting codes, such as Turbo codes, convolutional codes, or LDPC convolutional codes. Further, while the outer encoders are presently described as individually configured for each transmit antenna, no limitation is intended in this regard. A single outer encoder may be used for a plurality of transmit antennas, or the number of outer encoders may be greater than the number of transmit antennas.) The system also has interleavers (π a , π b ) for each of the streams A and B. Here, the modulation scheme is 2 h -QAM (i.e., h bits transmitted per symbol). 
     The receiver performs iterative detection (iterative APP (or Max-log APP) decoding) of MIMO signals, as described above. The LDPC codes are decoded using, for example, sum-product decoding. 
       FIG.  2    illustrates the frame configuration and describes the symbol order after interleaving. Here, (i a ,j a ) and (i b ,j b ) can be expressed as follows.
 
[Math. 16]
 
( i   a   ,j   a )=π a (Ω ia,ja   a )  (formula 16)
 
[Math. 17]
 
( i   b   ,j   b )=π b (Ω ib,jb   a )  (formula 17)
 
     Here, i a  and i b  represent the symbol order after interleaving, j a  and j b  represent the bit position in the modulation scheme (where j a ,j b =1, . . . , h), π a  and π b  represent the interleavers of streams A and B, and Ω a   ia,ja  and Ω b   ib,jb  represent the data order of streams A and B before interleaving. Note that  FIG.  2    illustrates a situation where i a =i b . 
     (Iterative Decoding) 
     The following describes, in detail, the sum-product decoding used in decoding the LDPC codes and the MIMO signal iterative detection algorithm, both used by the receiver. 
     Sum-Product Decoding 
     A two-dimensional M×N matrix H={H mn } is used as the check matrix for LDPC codes subject to decoding. For the set [1,N]={1, 2 . . . N}, the partial sets A(m) and B(n) are defined as follows.
 
[Math. 18]
 
 A ( m )≡{ n:H   mn =1}  (formula 18)
 
[Math. 19]
 
 B ( n )≡{ m:H   mn 1}  (formula 19)
 
     Here, A(m) signifies the set of column indices equal to 1 for row m of check matrix H, while B(n) signifies the set of row indices equal to 1 for row n of check matrix H. The sum-product decoding algorithm is as follows. 
     Step A-1 (Initialization): For all pairs (m,n) satisfying H mn =1, set the prior log ratio β mn =0. Set the loop variable (number of iterations) l sum =1, and set the maximum number of loops l sum,max . 
     Step A-2 (Processing): For all pairs (m,n) satisfying H mn =1 in the order m=1, 2, . . . M, update the extrinsic value log ratio α mn  using the following update formula. 
     
       
         
           
                             
             
               
                 
                   
                     [ 
                     
                       Math 
                       . 
                           
                       20 
                     
                     ] 
                   
                 
                 
                     
                 
               
             
           
         
       
       
         
           
             
               
                 
                   
                     α 
                     mn 
                   
                   = 
                   
                     
                       ( 
                       
                         
                           Π 
                           
                             
                               n 
                               ′ 
                             
                             ∈ 
                             
                               
                                 A 
                                 ⁡ 
                                 ( 
                                 m 
                                 ) 
                               
                               ⁢ 
                               \ 
                               ⁢ 
                               n 
                             
                           
                         
                         ⁢ 
                         
                           sign 
                           ( 
                           
                             
                               λ 
                               
                                 n 
                                 ′ 
                               
                             
                             + 
                             
                               β 
                               
                                 mn 
                                 ′ 
                               
                             
                           
                           ) 
                         
                       
                       ) 
                     
                     × 
                     
                       f 
                       ( 
                       
                         
                           ∑ 
                           
                             
                               n 
                               ′ 
                             
                             ∈ 
                             
                               
                                 A 
                                 ⁡ 
                                 ( 
                                 m 
                                 ) 
                               
                               ⁢ 
                               \ 
                               ⁢ 
                               n 
                             
                           
                         
                         
                           f 
                           ⁡ 
                           ( 
                           
                             
                               λ 
                               
                                 n 
                                 ′ 
                               
                             
                             + 
                             
                               β 
                               
                                 mn 
                                 ′ 
                               
                             
                           
                           ) 
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     20 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
                             
             
               
                 
                   
                     [ 
                     
                       Math 
                       . 
                           
                       21 
                     
                     ] 
                   
                 
                 
                     
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       sign 
                       ( 
                       x 
                       ) 
                     
                     ≡ 
                     
                       { 
                       
                         
                           
                             
                               1 
                                   
                             
                           
                           
                             
                               x 
                               ≥ 
                               0 
                             
                           
                         
                         
                           
                             
                               - 
                               1 
                             
                           
                           
                             
                               x 
                               &lt; 
                               0 
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     21 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
                             
             
               
                 
                   
                     [ 
                     
                       Math 
                       . 
                           
                       22 
                     
                     ] 
                   
                 
                 
                     
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       f 
                       ⁡ 
                       ( 
                       x 
                       ) 
                     
                     ≡ 
                     
                       ln 
                       ⁢ 
                       
                         
                           
                             exp 
                             ⁡ 
                             ( 
                             x 
                             ) 
                           
                           + 
                           1 
                         
                         
                           
                             exp 
                             ⁡ 
                             ( 
                             x 
                             ) 
                           
                           - 
                           1 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     22 
                   
                   ) 
                 
               
             
           
         
       
     
     where ƒ is the Gallager function. λ n  can then be computed as follows. 
     Step A-3 (Column Operations): For all pairs (m,n) satisfying H mn =1 in the order n=1, 2, . . . N, update the extrinsic value log ratio β mn  using the following update formula. 
             [     Math   .         23     ]                       β   mn     =       ∑       m   ′     ∈       B   ⁡   (   n   )     ⁢   \   ⁢   m           α       m   ′     ⁢   n                 (     formula   ⁢         23     )               
Step A-4 (Log-likelihood Ratio Calculation): For n∈[1,N], the log-likelihood ratio L n  is computed as follows.
 
             [     Math   .         24     ]                       L   n     =         ∑       m   ′     ∈       B   ⁡   (   n   )     ⁢   \   ⁢   m           α       m   ′     ⁢   n         +     λ   n               (     formula   ⁢         24     )               
Step A-5 (Iteration Count): If l sum &lt;l sum,max , then l sum  is incremented and the process returns to step A-2. Sum-product decoding ends when l sum =l sum,max .
 
     The above describes one iteration of sum-product decoding operations. Afterward, MIMO signal iterative detection is performed. The variables m, n, α mn , β mn , λ n , and L n  used in the above explanation of sum-product decoding operations are expressed as m a , n a , α a   mana , β a   mana , λ na , and L na  for stream A and as m b , n b , α b   mbnb , β b   mbnb , λ nb , and L nb  for stream B. 
     (MIMO Signal Iterative Detection) 
     The following describes the calculation of λ n  for MIMO signal iterative detection. 
     The following formula is derivable from Math. 1 (formula 1). 
     
       
         
           
             [ 
             
               Math 
               . 
                   
               25 
             
             ] 
           
         
       
       
         
           
             
               
                 
                   
                     
                       
                         
                           y 
                           ⁡ 
                           ( 
                           t 
                           ) 
                         
                         = 
                         
                           
                             ( 
                             
                               
                                 
                                   y 
                                   1 
                                 
                                 ( 
                                 t 
                                 ) 
                               
                               , 
                               
                                 
                                   y 
                                   2 
                                 
                                 ( 
                                 t 
                                 ) 
                               
                             
                             ) 
                           
                           T 
                         
                       
                     
                   
                   
                     
                       
                                   
                         
                           = 
                           
                             
                               
                                 
                                   H 
                                   22 
                                 
                                 ( 
                                 t 
                                 ) 
                               
                               ⁢ 
                               
                                 s 
                                 ⁡ 
                                 ( 
                                 t 
                                 ) 
                               
                             
                             + 
                             
                               n 
                               ⁡ 
                               ( 
                               t 
                               ) 
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     25 
                   
                   ) 
                 
               
             
           
         
       
     
     Given the frame configuration illustrated in  FIG.  2   , the following functions are derivable from Math. 16 (formula 16) and Math. 17 (formula 17).
 
[Math. 26]
 
 n   a =Ω ia,ja   a   (formula 26)
 
[Math. 27]
 
 n   b =Ω ib,jb   b   (formula 27)
 
     where n a ,n b ∈[1,N]. For iteration k of MIMO signal iterative detection, the variables λ na , L na , λ nb , and L nb  are expressed as λ k,na , L k,na , λ κ,nb , and L k,nb . 
     Step B-1 (Initial Detection; k=0) For initial wave detection, λ o,na  and 
     λ 0,nb  are calculated as follows. 
     For iterative APP decoding: 
                                   [     Math   .         28     ]     ⁢   
             λ     0   ,     n   X         =     ln   ⁢         Σ     U     0   ,     n   X     ,     +   1           ⁢   exp   ⁢     {       -     1     2   ⁢     σ   2           ⁢              y   ⁡   (     i   X     )     -         H   22     (     i   X     )     ⁢     s   ⁡   (     u   ⁡   (     i   X     )     )              2       }           Σ     U     0   ,     n   X     ,     -   1           ⁢     exp   (       -     1     2   ⁢     σ   2           ⁢              y   ⁡   (     i   X     )     -         H   22     (     i   X     )     ⁢     s   ⁡   (     u   ⁡   (     i   X     )     )              2       }                   (     formula   ⁢         28     )                   
For iterative Max-log APP decoding:
 
     
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                         
                       29 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                   
                     λ 
                     
                       0 
                       , 
                       
                         n 
                         X 
                       
                     
                   
                   = 
                   
                     
                       
                         max 
                         
                           U 
                           
                             0 
                             , 
                             
                               n 
                               X 
                             
                             , 
                             
                               + 
                               1 
                             
                           
                         
                       
                       
                         { 
                         
                           Ψ 
                           ⁡ 
                           ( 
                           
                             
                               u 
                               ⁡ 
                               ( 
                               
                                 i 
                                 X 
                               
                               ) 
                             
                             , 
                             
                               y 
                               ⁡ 
                               ( 
                               
                                 i 
                                 X 
                               
                               ) 
                             
                           
                           ) 
                         
                         } 
                       
                     
                     - 
                     
                       
                         max 
                         
                           U 
                           
                             0 
                             , 
                             
                               n 
                               X 
                             
                             , 
                             
                               - 
                               1 
                             
                           
                         
                       
                       
                         { 
                         
                           Ψ 
                           ⁡ 
                           ( 
                           
                             
                               u 
                               ⁡ 
                               ( 
                               
                                 i 
                                 X 
                               
                               ) 
                             
                             , 
                             
                               y 
                               ⁡ 
                               ( 
                               
                                 i 
                                 X 
                               
                               ) 
                             
                           
                           ) 
                         
                         } 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     29 
                   
                   ) 
                 
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     [ 
                     
                       Math 
                       . 
                           
                       30 
                     
                     ] 
                   
                 
               
               
                   
               
             
           
         
       
       
         
           
             
               
                 
                                   
                   
                     
                       Ψ 
                       ⁡ 
                       ( 
                       
                         
                           u 
                           ⁡ 
                           ( 
                           
                             i 
                             X 
                           
                           ) 
                         
                         , 
                         
                           y 
                           ⁡ 
                           ( 
                           
                             i 
                             X 
                           
                           ) 
                         
                       
                       ) 
                     
                     = 
                     
                       
                         - 
                         
                           1 
                           
                             2 
                             ⁢ 
                             
                               σ 
                               2 
                             
                           
                         
                       
                       ⁢ 
                       
                         
                            
                           
                             
                               y 
                               ⁡ 
                               ( 
                               
                                 i 
                                 X 
                               
                               ) 
                             
                             - 
                             
                               
                                 
                                   H 
                                   22 
                                 
                                 ( 
                                 
                                   i 
                                   X 
                                 
                                 ) 
                               
                               ⁢ 
                               
                                 s 
                                 ⁡ 
                                 ( 
                                 
                                   u 
                                   ⁡ 
                                   ( 
                                   
                                     i 
                                     X 
                                   
                                   ) 
                                 
                                 ) 
                               
                             
                           
                            
                         
                         2 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                         
                     30 
                   
                   ) 
                 
               
             
           
         
       
     
     where X=a,b. Next, the iteration count for the MIMO signal iterative detection is set to l mimo =0, with the maximum iteration count being l mimo,max . 
     Step B-2 (Iterative Detection; Iteration k): When the iteration count is k, Math. 11 (formula 11), Math. 13 (formula 13) through Math. 15 (formula 15), Math. 16 (formula 16), and Math. 17 (formula 17) can be expressed as Math. 31 (formula 31) through Math. 34 (formula 34), below. Note that (X,Y)=(a,b)(b,a). 
     For iterative APP decoding: 
                           ⁢     [     Math   .           ⁢   31     ]                               λ     k   ,     n   x         =         L       k   -   1     ,     Ω     iX   ,   jX     X         ⁡     (     u     Ω     iX   ,   jX     X       )       +     ln   ⁢         ∑       U     k   ,     n   x         +   1       ⁢           ⁢     exp   ⁢     {               -     1     2   ⁢           ⁢     σ   2           ⁢                    y   ⁢     (     i   X     )       -                   H   22     ⁡     (     i   X     )       ⁢     s   ⁡     (     u   ⁡     (     i   X     )       )                    2       +               ρ   ⁡     (     u     Ω     iX   ,   jX     X       )             }             ∑       U     k   ,     n   x         -   1       ⁢     exp   ⁢     {               -     1     2   ⁢           ⁢     σ   2           ⁢                    y   ⁢     (     i   X     )       -                   H   22     ⁡     (     i   X     )       ⁢     s   ⁡     (     u   ⁡     (     i   X     )       )                    2       +               ρ   ⁡     (     u     Ω     iX   ,   jX     X       )             }                       (     formula   ⁢           ⁢   31     )                       ⁢     [     Math   .           ⁢   32     ]                               ρ   ⁢     (     u     Ω     iX   ,   jX     X       )       =         ∑       γ   =   1       γ   ≠   jX       h     ⁢           ⁢                L       k   -   1     ,     Ω     iX   ,   γ     X         ⁡     (     u     Ω     iX   ,   γ     X       )       2          ⁢     (         u     Ω     iX   ,   γ     X       ⁢   sign   ⁢           ⁢     (       L       k   -   1     ,     Ω     iX   ,   γ     X         ⁡     (     u     Ω     iX   ,   γ     X       )       )       -   1     )         +       ∑     γ   =   1     h     ⁢                L       k   -   1     ,     Ω     iX   ,   γ     X         ⁡     (     u     Ω     iX   ,   γ     X       )       2          ⁢     (         u     Ω     iX   ,   γ     X       ⁢   sign   ⁢           ⁢     (       L       k   -   1     ,     Ω     iX   ,   γ     X         ⁡     (     u     Ω     iX   ,   γ     X       )       )       -   1     )                   (     formula   ⁢           ⁢   32     )               
For iterative Max-log APP decoding:
 
                           ⁢     [     Math   .           ⁢   33     ]                               λ     k   ,     n   x         =         L       k   -   1     ,     Ω     iX   ,   jX     X         ⁡     (     u     Ω     iX   ,   jX     X       )       +       max       U     k   ,     n   x         +   1       ⁢     {     Ψ   ⁡     (       u   ⁡     (     i   X     )       ,     y   ⁡     (     i   X     )       ,     ρ   ⁡     (     u     Ω     iX   ,   jX     X       )         )       }       -       max       U     k   ,     n   x         -   1       ⁢     {     Ψ   ⁡     (       u   ⁡     (     i   X     )       ,     y   ⁡     (     i   X     )       ,     ρ   ⁡     (     u     Ω     iX   ,   jX     X       )         )       }                 (     formula   ⁢           ⁢   33     )                       ⁢     [     Math   .           ⁢   34     ]                               Ψ   ⁡     (       u   ⁡     (     i   X     )       ,     y   ⁡     (     i   X     )       ,     ρ   ⁡     (     u     Ω     iX   ,   jX     X       )         )       =         -     1     2   ⁢           ⁢     σ   2           ⁢              y   ⁢     (     i   X     )       -         H   22     ⁡     (     i   X     )       ⁢     s   ⁡     (     u   ⁡     (     i   X     )       )                2       +     ρ   ⁡     (     u     Ω     iX   ,   jX     X       )                 (     formula   ⁢           ⁢   34     )               
Step B-3 (Iteration Count and Codeword Estimation) If l mimo &lt;l mimo,max , then l mimo  is incremented and the process returns to step B-2. When l mimo =l mimo,max , an estimated codeword is found, as follows.
 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     35 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       u 
                       ^ 
                     
                     
                       n 
                       
                         X 
                         = 
                       
                     
                   
                   ⁢ 
                   
                     { 
                     
                       
                         
                           1 
                         
                         
                           
                             
                               L 
                               
                                 
                                   l 
                                   mimo 
                                 
                                 , 
                                 
                                   n 
                                   X 
                                 
                               
                             
                             ≥ 
                             0 
                           
                         
                       
                       
                         
                           
                             - 
                             1 
                           
                         
                         
                           
                             
                               L 
                               
                                 
                                   l 
                                   mimo 
                                 
                                 , 
                                 
                                   n 
                                   X 
                                 
                               
                             
                             &lt; 
                             0 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     35 
                   
                   ) 
                 
               
             
           
         
       
     
     where X=a,b. 
       FIG.  3    shows a sample configuration of a transmission device  300  pertaining to the present Embodiment. An encoder  302 A takes information (data)  301 A and a frame configuration signal  313  as input (which includes the error-correction method, coding rate, block length, and other information used by the encoder  302 A in error-correction coding of the data, such that the method designated by the frame configuration signal  313  is used. The error-correction method may be switched). In accordance with the frame configuration signal  313 , the encoder  302 A performs error-correction coding, such as convolutional encoding, LDPC encoding, turbo encoding or similar, and outputs encoded data  303 A. 
     An interleaver  304 A takes the encoded data  303 A and the frame configuration signal  313  as input, performs interleaving, i.e., rearranges the order thereof, and then outputs interleaved data  305 A. (Depending on the frame configuration signal  313 , the interleaving method may be switched.) 
     A mapper  306 A takes the interleaved data  305 A and the frame configuration signal  313  as input and performs modulation, such as (Quadrature Phase Shift Keying), 16-QAM (16-Quadrature Amplitude Modulation), or 64-QAM (64-Quadrature Amplitude Modulation) thereon, then outputs a baseband signal  307 A. (Depending on the frame configuration signal  313 , the modulation scheme may be switched.) 
       FIGS.  19 A and  19 B  illustrate an example of a QPSK modulation mapping method for a baseband signal made up of an in-phase component I and a quadrature component Q in the I-Q plane. For example, as shown in  FIG.  19 A , when the input data are 00, then the output is I=1.0, Q=1.0. Similarly, when the input data are 01, the output is I=−1.0, Q=1.0, and so on.  FIG.  19 B  illustrates an example of a QPSK modulation mapping method in the I-Q plane differing from  FIG.  19 A  in that the signal points of  FIG.  19 A  have been rotated about the origin to obtain the signal points of  FIG.  19 B . Non-Patent Literature 9 and Non-Patent Literature 10 describe such a constellation rotation method. Alternatively, the Cyclic Q Delay described in Non-Patent Literature 9 and Non-Patent Literature 10 may also be adopted. An alternate example, distinct from  FIGS.  19 A and  19 B , is shown in  FIGS.  20 A and  20 B , which illustrate signal point distribution for 16-QAM in the I-Q plane. The example of  FIG.  20 A  corresponds to  FIG.  19 A , while that of  FIG.  20 B  corresponds to  FIG.  19 B . 
     An encoder  302 B takes information (data)  301 B and the frame configuration signal  313  as input (which includes the error-correction method, coding rate, block length, and other information used by the encoder  302 B in error-correction coding of the data, such that the method designated by the frame configuration signal  313  is used. The error-correction method may be switched). In accordance with the frame configuration signal  313 , the encoder  302 B performs error-correction coding, such as convolutional encoding, LDPC encoding, turbo encoding or similar, and outputs encoded data  303 B. 
     An interleaver  304 B takes the encoded data  303 B and the frame configuration signal  313  as input, performs interleaving, i.e., rearranges the order thereof, and outputs interleaved data  305 B. (Depending on the frame configuration signal  313 , the interleaving method may be switched.) 
     A mapper  306 B takes the interleaved data  305 B and the frame configuration signal  313  as input and performs modulation, such as QPSK, 16-QAM, or 64-QAM thereon, then outputs a baseband signal  307 B. (Depending on the frame configuration signal  313 , the modulation scheme may be switched.) 
     A signal processing method information generator  314  takes the frame configuration signal  313  as input and accordingly outputs signal processing method information  315 . The signal processing method information  315  designates the fixed precoding matrix to be used, and includes information on the pattern of phase changes used for changing the phase. 
     A weighting unit  308 A takes baseband signal  307 A, baseband signal  307 B, and the signal processing method information  315  as input and, in accordance with the signal processing method information  315 , performs weighting on the baseband signals  307 A and  307 B, then outputs a weighted signal  309 A. The weighting method is described in detail, later. 
     A wireless unit  310 A takes weighted signal  309 A as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal  311 A. Transmit signal  311 A is then output as radio waves by an antenna  312 A. 
     A weighting unit  308 B takes baseband signal  307 A, baseband signal  307 B, and the signal processing method information  315  as input and, in accordance with the signal processing method information  315 , performs weighting on the baseband signals  307 A and  307 B, then outputs weighted signal  316 B. 
       FIG.  21    illustrates the configuration of the weighting units  308 A and  308 B. The area of  FIG.  21    enclosed in the dashed line represents one of the weighting units. Baseband signal  307 A is multiplied by w 11  to obtain w 11 ·s 1 ( t ), and multiplied by w 21  to obtain w 21 ·s 1 ( t ). Similarly, baseband signal  307 B is multiplied by w 12  to obtain w 12 ·s 2 ( t ), and multiplied by w 22  to obtain w 22 ·s 2 ( t ). Next, z 1 ( t )=w 11 ·s 1 ( t )+w 12 ·s 2 ( t ) and z 2 ( t )=w 21 ·s 1 ( t )+w 22 ·s 22 ( t ) are obtained. Here, as explained in Embodiment 1, s 1 ( t ) and s 2 ( t ) are baseband signals modulated according to a modulation scheme such as BPSK (Binary Phase Shift Keying), QPSK, 8-PSK (8-Phase Shift Keying), 16-QAM, 32-QAM (32-Quadrature Amplitude Modulation), 64-QAM, 256-QAM 16-APSK (16-Amplitude Phase Shift Keying) and so on. 
     Both weighting units perform weighting using a fixed precoding matrix. The precoding matrix uses, for example, the method of Math. 36 (formula 36), and satisfies the conditions of Math. 37 (formula 37) or Math. 38 (formula 38), all found below. However, this is only an example. The value of α is not restricted to Math. 37 (formula 37) and Math. 38 (formula 38), and may take on other values, e.g., α=1. 
     Here, the precoding matrix is 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     36 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     ( 
                     
                       
                         
                           
                             w 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             11 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             12 
                           
                         
                       
                       
                         
                           
                             w 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             21 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             2 
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     
                       1 
                       
                         
                           
                             α 
                             2 
                           
                           + 
                           1 
                         
                       
                     
                     ⁢ 
                     
                       ( 
                       
                         
                           
                             
                               e 
                               
                                 j 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 0 
                               
                             
                           
                           
                             
                               α 
                               × 
                               
                                 e 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   0 
                                 
                               
                             
                           
                         
                         
                           
                             
                               α 
                               × 
                               
                                 e 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   0 
                                 
                               
                             
                           
                           
                             
                               e 
                               
                                 j 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 π 
                               
                             
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formul 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     a 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     36 
                   
                   ) 
                 
               
             
           
         
       
     
     In Math. 36 (formula 36), above, α is given by: 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     37 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   α 
                   = 
                   
                     
                       
                         2 
                       
                       + 
                       4 
                     
                     
                       
                         2 
                       
                       + 
                       2 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formul 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     a 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     37 
                   
                   ) 
                 
               
             
           
         
       
     
     Alternatively, in Math. 36 (formula 36), above, α may be given by: 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     38 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   α 
                   = 
                   
                     
                       
                         2 
                       
                       + 
                       3 
                       + 
                       
                         5 
                       
                     
                     
                       
                         2 
                       
                       + 
                       3 
                       - 
                       
                         5 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formul 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     a 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     38 
                   
                   ) 
                 
               
             
           
         
       
     
     The precoding matrix is not restricted to that of Math. 36 (formula 36), but may also be as indicated by Math. 39 (formula 39). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     39 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     ( 
                     
                       
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             2 
                           
                         
                       
                       
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             2 
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     ( 
                     
                       
                         
                           a 
                         
                         
                           b 
                         
                       
                       
                         
                           c 
                         
                         
                           d 
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     39 
                   
                   ) 
                 
               
             
           
         
       
     
     In Math. 39 (formula 39), let α=Ae jδ11 , b=Be jδ12 , c=Ce jδ21 , and d=De jδ22 . Further, one of a, b, c, and d may be equal to zero. For example, the following configurations are possible: (1) a may be zero while b, c, and d are non-zero, (2) b may be zero while a, c, and d are non-zero, (3) c may be zero while a, b, and d are non-zero, or (4) d may be zero while a, b, and c are non-zero. 
     When any of the modulation scheme, error-correcting codes, and the coding rate thereof are changed, the precoding matrix may also be set, changed, and fixed for use. 
     A phase changer  317 B takes weighted signal  316 B and the signal processing method information  315  as input, then regularly changes the phase of the signal  316 B for output. This regular change is a change of phase performed according to a predetermined phase changing pattern having a predetermined period (cycle) (e.g., every n symbols (n being an integer, n≥1) or at a predetermined interval). The details of the phase changing pattern are explained below, in Embodiment 4. 
     Wireless unit  310 B takes post-phase change signal  309 B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal  311 B. Transmit signal  311 B is then output as radio waves by an antenna  312 B. 
       FIG.  4    illustrates a sample configuration of a transmission device  400  that differs from that of  FIG.  3   . The points of difference of  FIG.  4    from  FIG.  3    are described next. 
     An encoder  402  takes information (data)  401  and the frame configuration signal  313  as input, and, in accordance with the frame configuration signal  313 , performs error-correction coding and outputs encoded data  402 . 
     A distributor  404  takes the encoded data  403  as input, performs distribution thereof, and outputs data  405 A and data  405 B. Although  FIG.  4    illustrates only one encoder, the number of encoders is not limited as such. The present invention may also be realized using m encoders (m being an integer, m≥1) such that the distributor divides the encoded data created by each encoder into two groups for distribution. 
       FIG.  5    illustrates an example of a frame configuration in the time domain for a transmission device according to the present Embodiment. Symbol  500 _ 1  is a symbol for notifying the reception device of the transmission scheme. For example, symbol  500 _ 1  conveys information such as the error-correction method used for transmitting data symbols, the coding rate thereof, and the modulation scheme used for transmitting data symbols. 
     Symbol  501 _ 1  is for estimating channel fluctuations for modulated signal z 1 ( t ) (where t is time) transmitted by the transmission device. Symbol  502 _ 1  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u (in the time domain). Symbol  503 _ 1  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u+1. 
     Symbol  501 _ 2  is for estimating channel fluctuations for modulated signal z 2 ( t ) (where t is time) transmitted by the transmission device. Symbol  502 _ 2  is a data symbol transmitted by modulated signal z 2 ( t ) as symbol number u. Symbol  503 _ 2  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u+1. 
     Here, the symbols of z 1 ( t ) and of z 2 ( t ) having the same timestamp (identical timing) are transmitted from the transmit antenna using the same (shared/common) frequency. 
     The following describes the relationships between the modulated signals z 1 ( t ) and z 2 ( t ) transmitted by the transmission device and the received signals r 1 ( t ) and r 2 ( t ) received by the reception device. 
     In  FIGS.  5 ,  504    # 1  and  504  # 2  indicate transmit antennas of the transmission device, while  505  # 1  and  505  # 2  indicate receive antennas of the reception device. The transmission device transmits modulated signal z 1 ( t ) from transmit antenna  504  # 1  and transmits modulated signal z 2 ( t ) from transmit antenna  504  # 2 . Here, modulated signals z 1 ( t ) and z 2 ( t ) are assumed to occupy the same (shared/common) frequency (bandwidth). The channel fluctuations in the transmit antennas of the transmission device and the antennas of the reception device are h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ), respectively. Assuming that receive antenna  505  # 1  of the reception device receives received signal r 1 ( t ) and that receive antenna  505  # 2  of the reception device receives received signal r 2 ( t ), the following relationship holds. 
     
       
         
           
             
               
                 
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       FIG.  6    pertains to the weighting method (precoding method) and the phase changing method of the present Embodiment. A weighting unit  600  is a combined version of the weighting units  308 A and  308 B from  FIG.  3   . As shown, stream s 1 ( t ) and stream s 2 ( t ) correspond to the baseband signals  307 A and  307 B of  FIG.  3   . That is, the streams s 1 ( t ) and s 2 ( t ) are baseband signals made up of an in-phase component I and a quadrature component Q conforming to mapping by a modulation scheme such as QPSK, 16-QAM, and 64-QAM. As indicated by the frame configuration of  FIG.  6   , stream s 1 ( t ) is represented as s 1 ( u ) at symbol number u, as s 1 ( u +1) at symbol number u+1, and so forth. Similarly, stream s 2 ( t ) is represented as s 2 ( u ) at symbol number u, as s 2 ( u +1) at symbol number u+1, and so forth. The weighting unit  600  takes the baseband signals  307 A (s 1 ( t )) and  307 B (s 2 ( t )) as well as the signal processing method information  315  from  FIG.  3    as input, performs weighting in accordance with the signal processing method information  315 , and outputs the weighted signals  309 A (z 1 ( t )) and  316 B(z 2 ′( t )) from  FIG.  3   . The phase changer  317 B changes the phase of weighted signal  316 B(z 2 ′( t )) and outputs post-phase change signal  309 B(z 2 ( t )). 
     Here, given vector W 1 =(w 11 ,w 12 ) from the first row of the fixed precoding matrix F, z 1 ( t ) is expressible as Math. 41 (formula 41), below.
 
[Math. 41]
 
 z 1= W 1×( s 1( t ), s 2( t )) T   (formula 41)
 
     Similarly, given vector W 2 =(w 21 ,w 22 ) from the second row of the fixed precoding matrix F, and letting the phase changing formula applied by the phase changer by y(t), then z 2 ( t ) is expressible as Math. 42 (formula 42), below.
 
[Math. 42]
 
 z 2( t )= y ( t )× W 2×( s 1( t ), s 2( t )) T   (formula 42)
 
     Here, y(t) is a phase changing formula obeying a predetermined method. For example, given a period (cycle) of four and timestamp u, the phase changing formula may be expressed as Math. 43 (formula 43), below.
 
[Math. 43]
 
 y ( u )= e   j0   (formula 43)
 
     Similarly, the phase changing formula for timestamp u+1 may be, for example, as given by Math. 44 (formula 44). 
     
       
         
           
             
               
                 
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     That is, the phase changing formula for timestamp u+k generalizes to Math. 45 (formula 45). 
     
       
         
           
             
               
                 
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     Note that Math. 43 (formula 43) through Math. 45 (formula 45) are given only as an example of a regular change of phase. 
     The regular change of phase is not restricted to a period (cycle) of four. Improved reception capabilities (the error-correction capabilities, to be exact) may potentially be promoted in the reception device by increasing the period (cycle) number (this does not mean that a greater period (cycle) is better, though avoiding small numbers such as two is likely ideal). 
     Furthermore, although Math. 43 (formula 43) through Math. 45 (formula 45), above, represent a configuration in which a change in phase is carried out through rotation by consecutive predetermined phases (in the above formula, every π/2), the change in phase need not be rotation by a constant amount, but may also be random. For example, in accordance with the predetermined period (cycle) of y(t), the phase may be changed through sequential multiplication as shown in Math. 46 (formula 46) and Math. 47 (formula 47). The key point of the regular change of phase is that the phase of the modulated signal is regularly changed. The phase changing degree variance rate is preferably as even as possible, such as from −π radians to π radians. However, given that this concerns a distribution, random variance is also possible. 
     
       
         
           
             
               
                 
                   
                       
                   
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     As such, the weighting unit  600  of  FIG.  6    performs precoding using fixed, predetermined precoding weights, and the phase changer  317 B changes the phase of the signal input thereto while regularly varying the phase changing degree. 
     When a specialized precoding matrix is used in the LOS environment, the reception quality is likely to improve tremendously. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may greatly differ from the specialized precoding matrix, upon reception. The LOS environment has certain rules. Thus, data reception quality is tremendously improved through a regular change of transmit signal phase that obeys those rules. The present invention offers a signal processing method for improving the LOS environment. 
       FIG.  7    illustrates a sample configuration of a reception device  700  pertaining to the present embodiment. Wireless unit  703 _X receives, as input, received signal  702 _X received by antenna  701 _X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal  704 _X. 
     Channel fluctuation estimator  705 _ 1  for modulated signal z 1  transmitted by the transmission device takes baseband signal  704 _X as input, extracts reference symbol  501 _ 1  for channel estimation from  FIG.  5   , estimates the value of h 11  from Math. 40 (formula 40), and outputs channel estimation signal  706 _ 1 . 
     Channel fluctuation estimator  705 _ 2  for modulated signal z 2  transmitted by the transmission device takes baseband signal  704 _X as input, extracts reference symbol  502 _ 2  for channel estimation from  FIG.  5   , estimates the value of h 12  from Math. 40 (formula 40), and outputs channel estimation signal  706 _ 1 . 
     Wireless unit  703 _Y receives, as input, received signal  702 _Y received by antenna  701 _Y, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal  704 _Y. 
     Channel fluctuation estimator  707 _ 1  for modulated signal z 1  transmitted by the transmission device takes baseband signal  704 _Y as input, extracts reference symbol  501 _ 1  for channel estimation from  FIG.  5   , estimates the value of h 11  from Math. 40 (formula 40), and outputs channel estimation signal  708 _ 1 . 
     Channel fluctuation estimator  707 _ 2  for modulated signal z 2  transmitted by the transmission device takes baseband signal  704 _Y as input, extracts reference symbol  502 _ 2  for channel estimation from  FIG.  5   , estimates the value of h 11  from Math. 40 (formula 40), and outputs channel estimation signal  708 _ 2 . 
     A control information decoder  709  receives baseband signal  704 _X and baseband signal  704 _Y as input, detects symbol  500 _ 1  that indicates the transmission scheme from  FIG.  5   , and outputs a transmission method information signal  710  for the transmission device. 
     A signal processor  711  takes the baseband signals  704 _X and  704 _Y, the channel estimation signals  706 _ 1 ,  706 _ 2 ,  708 _ 1 , and  708 _ 2 , and the transmission method information signal  710  as input, performs detection and decoding, and then outputs received data  712 _ 1  and  712 _ 2 . 
     Next, the operations of the signal processor  711  from  FIG.  7    are described in detail.  FIG.  8    illustrates a sample configuration of the signal processor  711  pertaining to the present embodiment. As shown, the signal processor  711  is primarily made up of an inner MIMO detector, a soft-in/soft-out decoder, and a coefficient generator. Non-Patent Literature 2 and Non-Patent Literature 3 describe the method of iterative decoding with this structure. The MIMO system described in Non-Patent Literature 2 and Non-Patent Literature 3 is a spatial multiplexing MIMO system, while the present Embodiment differs from Non-Patent Literature 2 and Non-Patent Literature 3 in describing a MIMO system that regularly changes the phase over time, while using the precoding matrix. Taking the (channel) matrix H(t) of Math. 36 (formula 36), then by letting the precoding weight matrix from  FIG.  6    be F (here, a fixed precoding matrix remaining unchanged for a given received signal) and letting the phase changing formula used by the phase changer from  FIG.  6    be Y(t) (here, Y(t) changes over time t), then the receive vector R(t)=(r 1 ( t ),r 2 ( t )) T  and the stream vector S(t)=(s 1 ( t ),s 2 ( t )) T  the following function is derived: 
     
       
         
           
             
               
                 
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                     . 
                     
                         
                     
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                     48 
                   
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     Here, the reception device may use the decoding methods of Non-Patent Literature 2 and 3 on R(t) by computing H(t)×Y(t)×F. 
     Accordingly, the coefficient generator  819  from  FIG.  8    takes a transmission method information signal  818  (corresponding to  710  from  FIG.  7   ) indicated by the transmission device (information for specifying the fixed precoding matrix in use and the phase changing pattern used when the phase is changed) and outputs a signal processing method information signal  820 . 
     The inner MIMO detector  803  takes the signal processing method information signal  820  as input and performs iterative detection and decoding using the signal and the relationship thereof to Math. 48 (formula 48). The operations thereof are described below. 
     The processing unit illustrated in  FIG.  8    must use a processing method, as is illustrated in  FIG.  10   , to perform iterative decoding (iterative detection). First, detection of one codeword (or one frame) of modulated signal (stream) s 1  and of one codeword (or one frame) of modulated signal (stream) s 2  are performed. As a result, the soft-in/soft-out decoder obtains the log-likelihood ratio of each bit of the codeword (or frame) of modulated signal (stream) s 1  and of the codeword (or frame) of modulated signal (stream) s 2 . Next, the log-likelihood ratio is used to perform a second round of detection and decoding. These operations (referred to as iterative decoding (iterative detection)) are performed multiple times. The following explanations centre on the creation method of the log-likelihood ratio of a symbol at a specific time within one frame. 
     In  FIG.  8   , a memory  815  takes baseband signal  801 X (corresponding to baseband signal  704 _X from  FIG.  7   ), channel estimation signal group  802 X (corresponding to channel estimation signals  706 _ 1  and  706 _ 2  from  FIG.  7   ), baseband signal  801 Y (corresponding to baseband signal  704 _Y from  FIG.  7   ), and channel estimation signal group  802 Y (corresponding to channel estimation signals  708 _ 1  and  708 _ 2  from  FIG.  7   ) as input, executes (computes) H(t)×Y(t)×F from Math. 48 (formula 48) in order to perform iterative decoding (iterative detection), and stores the resulting matrix as a transformed channel signal group. The memory  815  then outputs the above-described signals as needed, specifically as baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, and transformed channel estimation signal group  817 Y. 
     Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection). 
     (Initial Detection) 
     The inner MIMO detector  803  takes baseband signal  801 X, channel estimation signal group  802 X, baseband signal  801 Y, and channel estimation signal group  802 Y as input. Here, the modulation scheme for modulated signal (stream) s 1  and modulated signal (stream) s 2  is described as 16-QAM. 
     The inner MIMO detector  803  first computes H(t)×Y(t)×F from the channel estimation signal groups  802 X and  802 Y, thus calculating a candidate signal point corresponding to baseband signal  801 X.  FIG.  11    represents such a calculation. In  FIG.  11   , each black dot is a candidate signal point in the I-Q plane. Given that the modulation scheme is 16-QAM, 256 candidate signal points exist. (However,  FIG.  11    is only a representation and does not indicate all 256 candidate signal points.) Letting the four bits transmitted in modulated signal s 1  be b 0 , b 1 , b 2 , and b 3  and the four bits transmitted in modulated signal s 2  be b 4 , b 5 , b 6 , and b 7 , candidate signal points corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) are found in  FIG.  11   . The Euclidean squared distance between each candidate signal point and each received signal point  1101  (corresponding to baseband signal  801 X) is then computed. The Euclidian squared distance between each point is divided by the noise variance σ 2 . Accordingly, E X (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is calculated. That is, the Euclidian squared distance between a candidate signal point corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) and a received signal point is divided by the noise variance. Here, each of the baseband signals and the modulated signals s 1  and s 2  is a complex signal. 
     Similarly, the inner MIMO detector  803  computes H(t)×Y(t)×F from the channel estimation signal groups  802 X and  802 Y, calculates candidate signal points corresponding to baseband signal  801 Y, computes the Euclidean squared distance between each of the candidate signal points and the received signal points (corresponding to baseband signal  801 Y), and divides the Euclidean squared distance by the noise variance σ 2 . Accordingly, E Y (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is calculated. That is, E Y  is the Euclidian squared distance between a candidate signal point corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) and a received signal point, divided by the noise variance. 
     Next, E X (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 )+E Y (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 )=E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is computed. 
     The inner MIMO detector  803  outputs E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) as the signal  804 . 
     The log-likelihood calculator  805 A takes the signal  804  as input, calculates the log-likelihood of bits b 0 , b 1 , b 2 , and b 3 , and outputs the log-likelihood signal  806 A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 and the log-likelihood of a bit being 0. The calculation method is as shown in Math. 28 (formula 28), Math. 29 (formula 29), and Math. 30 (formula 30), and the details thereof are given by Non-Patent Literature 2 and 3. 
     Similarly, log-likelihood calculator  805 B takes the signal  804  as input, calculates the log-likelihood of bits b 4 , b 5 , b 6 , and b 7 , and outputs log-likelihood signal  806 B. 
     A deinterleaver ( 807 A) takes log-likelihood signal  806 A as input, performs deinterleaving corresponding to that of the interleaver (the interleaver ( 304 A) from  FIG.  3   ), and outputs deinterleaved log-likelihood signal  808 A. 
     Similarly, a deinterleaver ( 807 B) takes log-likelihood signal  806 B as input, performs deinterleaving corresponding to that of the interleaver (the interleaver ( 304 B) from  FIG.  3   ), and outputs deinterleaved log-likelihood signal  808 B. 
     Log-likelihood ratio calculator  809 A takes deinterleaved log-likelihood signal  808 A as input, calculates the log-likelihood ratio of the bits encoded by encoder  302 A from  FIG.  3   , and outputs log-likelihood ratio signal  810 A. 
     Similarly, log-likelihood ratio calculator  809 B takes deinterleaved log-likelihood signal  808 B as input, calculates the log-likelihood ratio of the bits encoded by encoder  302 B from  FIG.  3   , and outputs log-likelihood ratio signal  810 B. 
     Soft-in/soft-out decoder  811 A takes log-likelihood ratio signal  810 A as input, performs decoding, and outputs a decoded log-likelihood ratio  812 A. 
     Similarly, soft-in/soft-out decoder  811 B takes log-likelihood ratio signal  810 B as input, performs decoding, and outputs decoded log-likelihood ratio  812 B. 
     (Iterative Decoding (Iterative Detection), k Iterations) 
     The interleaver ( 813 A) takes the k−1th decoded log-likelihood ratio  812 A decoded by the soft-in/soft-out decoder as input, performs interleaving, and outputs an interleaved log-likelihood ratio  814 A. Here, the interleaving pattern used by the interleaver ( 813 A) is identical to that of the interleaver ( 304 A) from  FIG.  3   . 
     Another interleaver ( 813 B) takes the k−1th decoded log-likelihood ratio  812 B decoded by the soft-in/soft-out decoder as input, performs interleaving, and outputs interleaved log-likelihood ratio  814 B. Here, the interleaving pattern used by the interleaver ( 813 B) is identical to that of the other interleaver ( 304 B) from  FIG.  3   . 
     The inner MIMO detector  803  takes baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, transformed channel estimation signal group  817 Y, interleaved log-likelihood ratio  814 A, and interleaved log-likelihood ratio  814 B as input. Here, baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, and transformed channel estimation signal group  817 Y are used instead of baseband signal  801 X, channel estimation signal group  802 X, baseband signal  801 Y, and channel estimation signal group  802 Y because the latter cause delays due to the iterative decoding. 
     The iterative decoding operations of the inner MIMO detector  803  differ from the initial detection operations thereof in that the interleaved log-likelihood ratios  814 A and  814 B are used in signal processing for the former. The inner MIMO detector  803  first calculates E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) in the same manner as for initial detection. In addition, the coefficients corresponding to Math. 11 (formula 11) and Math. 32 (formula 32) are computed from the interleaved log-likelihood ratios  814 A and  814 B. The value of E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is corrected using the coefficients so calculated to obtain E′(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ), which is output as the signal  804 . 
     The log-likelihood calculator  805 A takes the signal  804  as input, calculates the log-likelihood of bits b 0 , b 1 , b 2 , and b 3 , and outputs the log-likelihood signal  806 A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 and the log-likelihood of a bit being 0. The calculation method is as shown in Math. 31 (formula 31) through Math. 35 (formula 35), and the details are given by Non-Patent Literature 2 and 3. 
     Similarly, log-likelihood calculator  805 B takes the signal  804  as input, calculates the log-likelihood of bits b 4 , b 5 , b 6 , and b 7 , and outputs log-likelihood signal  806 B. Operations performed by the deinterleaver onwards are similar to those performed for initial detection. 
     While  FIG.  8    illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary as good reception improvements are obtainable by iterative detection alone. As long as the components needed for iterative detection are present, the configuration need not include the interleavers  813 A and  813 B. In such a case, the inner MIMO detector  803  does not perform iterative detection. 
     The key point for the present Embodiment is the calculation of H(t)×Y(t)×F. As shown in Non-Patent Literature 5 and the like, QR decomposition may also be used to perform initial detection and iterative detection. 
     Also, as indicated by Non-Patent Literature 11, MMSE (Minimum Mean-Square Error) and ZF (Zero-Forcing) linear operations may be performed based on H(t)×Y(t)×F when performing initial detection. 
       FIG.  9    illustrates the configuration of a signal processor, unlike that of  FIG.  8   , that serves as the signal processor for modulated signals transmitted by the transmission device from  FIG.  4   . The point of difference from  FIG.  8    is the number of soft-in/soft-out decoders. A soft-in/soft-out decoder  901  takes the log-likelihood ratio signals  810 A and  810 B as input, performs decoding, and outputs a decoded log-likelihood ratio  902 . A distributor  903  takes the decoded log-likelihood ratio  902  as input for distribution. Otherwise, the operations are identical to those explained for  FIG.  8   . 
     As described above, when a transmission device according to the present Embodiment using a MIMO system transmits a plurality of modulated signals from a plurality of antennas, changing the phase over time while multiplying by the precoding matrix so as to regularly change the phase results in improvements to data reception quality for a reception device in a LOS environment, where direct waves are dominant, compared to a conventional spatial multiplexing MIMO system. 
     In the present Embodiment, and particularly in the configuration of the reception device, the number of antennas is limited and explanations are given accordingly. However, the Embodiment may also be applied to a greater number of antennas. In other words, the number of antennas in the reception device does not affect the operations or advantageous effects of the present Embodiment. 
     Also, although LDPC codes are described as a particular example, the present Embodiment is not limited in this manner, Furthermore, the decoding method is not limited to the sum-product decoding example given for the soft-in/soft-out decoder. Other soft-in/soft-out decoding methods, such as the BCJR algorithm, SOVA, and the Max-Log-Map algorithm may also be used. Details are provided in Non-Patent Literature 6. 
     In addition, although the present Embodiment is described using a single-carrier method, no limitation is intended in this regard. The present Embodiment is also applicable to multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA (Single Carrier Frequency-Division Multiple Access), SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, and so on) or symbols transmitting control information, may be arranged within the frame in any manner. 
     The following describes an example in which OFDM is used as a multi-carrier method. 
       FIG.  12    illustrates the configuration of a transmission device using OFDM. In  FIG.  12   , components operating in the manner described for  FIG.  3    use identical reference numbers. 
     An OFDM-related processor  1201 A takes weighted signal  309 A as input, performs OFDM-related processing thereon, and outputs transmit signal  1202 A. Similarly, OFDM-related processor  1201 B takes post-phase change signal  309 B as input, performs OFDM-related processing thereon, and outputs transmit signal  1202 B. 
       FIG.  13    illustrates a sample configuration of the OFDM-related processors  1201 A and  1201 B and onward from  FIG.  12   . Components  1301 A through  1310 A belong between  1201 A and  312 A from  FIG.  12   , while components  1301 B through  1310 B belong between  1201 B and  312 B. 
     Serial-to-parallel converter  1302 A performs serial-to-parallel conversion on weighted signal  1301 A (corresponding to weighted signal  309 A from  FIG.  12   ) and outputs parallel signal  1303 A. 
     Reorderer  1304 A takes parallel signal  1303 A as input, performs reordering thereof, and outputs reordered signal  1305 A. Reordering is described in detail later. 
     IFFT (Inverse Fast Fourier Transform) unit  1306 A takes reordered signal  1305 A as input, applies an IFFT thereto, and outputs post-IFFT signal  1307 A. 
     Wireless unit  1308 A takes post-IFFT signal  1307 A as input, performs processing such as frequency conversion and amplification, thereon, and outputs modulated signal  1309 A. Modulated signal  1309 A is then output as radio waves by antenna  1310 A. 
     Serial-to-parallel converter  1302 B performs serial-to-parallel conversion on weighted signal  1301 B (corresponding to post-phase change  309 B from  FIG.  12   ) and outputs parallel signal  1303 B. 
     Reorderer  1304 B takes parallel signal  1303 B as input, performs reordering thereof, and outputs reordered signal  1305 B. Reordering is described in detail later. 
     IFFT unit  1306 B takes reordered signal  1305 B as input, applies an IFFT thereto, and outputs post-IFFT signal  1307 B. 
     Wireless unit  1308 B takes post-IFFT signal  1307 B as input, performs processing such as frequency conversion and amplification thereon, and outputs modulated signal  1309 B. Modulated signal  1309 B is then output as radio waves by antenna  1310 A. 
     The transmission device from  FIG.  3    does not use a multi-carrier transmission method. Thus, as shown in  FIG.  6   , a change of phase is performed to achieve a period (cycle) of four and the post-phase change symbols are arranged in the time domain. As shown in  FIG.  12   , when multi-carrier transmission, such as OFDM, is used, then, naturally, precoded post-phase change symbols may be arranged with respect to the time domain as in  FIG.  3   , and this applies to each (sub-)carrier. However, for multi-carrier transmission, the arrangement may also be in the frequency domain, or in both the frequency domain and the time domain. The following describes these arrangements. 
       FIGS.  14 A and  14 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13   . The frequency axes are made up of (sub-)carriers  0  through  9 . The modulated signals z 1  and z 2  share common timestamps (timing) and use a common frequency band.  FIG.  14 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  14 B  illustrates a reordering method for the symbols of modulated signal z 2 . With respect to the symbols of weighted signal  1301 A input to serial-to-parallel converter  1302 A, the assigned ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, #0, #1, #2, and #3 are equivalent to one period (cycle). Similarly, #4n, #4n+1, #4n+2, and #4n+3 (n being a non-zero positive integer) are also equivalent to one period (cycle). 
     As shown in  FIG.  14 A , symbols #0, #1, #2, #3, and so on are arranged in order, beginning at carrier  0 . Symbols #0 through #9 are given timestamp $ 1 , followed by symbols #10 through #19 which are given timestamp #2, and so on in a regular arrangement. Here, modulated signals z 1  and z 2  are complex signals. 
     Similarly, with respect to the symbols of weighted signal  1301 B input to serial-to-parallel converter  1302 B, the assigned ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, a different change in phase is applied to each of #0, #1, #2, and #3, which are equivalent to one period (cycle). Similarly, a different change in phase is applied to each of #4n, #4n+1, #4n+2, and #4n+3 (n being a non-zero positive integer), which are also equivalent to one period (cycle). 
     As shown in  FIG.  14 B , symbols #0, #1, #2, #3, and so on are arranged in order, beginning at carrier  0 . Symbols #0 through #9 are given timestamp $ 1 , followed by symbols #10 through #19 which are given timestamp $ 2 , and so on in a regular arrangement. 
     The symbol group  1402  shown in  FIG.  14 B  corresponds to one period (cycle) of symbols when the phase changing method of  FIG.  6    is used. Symbol #0 is the symbol obtained by using the phase at timestamp u in  FIG.  6   , symbol #1 is the symbol obtained by using the phase at timestamp u+1 in  FIG.  6   , symbol #2 is the symbol obtained by using the phase at timestamp u+2 in  FIG.  6   , and symbol #3 is the symbol obtained by using the phase at timestamp u+3 in  FIG.  6   . Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at timestamp u in  FIG.  6    when x mod 4 equals 0 (i.e., when the remainder of x divided by 4 is 0, mod being the modulo operator), symbol #x is the symbol obtained by using the phase at timestamp u+1 in  FIG.  6    when x mod 4 equals 1, symbol #x is the symbol obtained by using the phase at timestamp u+2 in  FIG.  6    when x mod 4 equals 2, and symbol #x is the symbol obtained by using the phase at timestamp u+3 in  FIG.  6    when x mod 4 equals 3. 
     In the present Embodiment, modulated signal z 1  shown in  FIG.  14 A  has not undergone a change of phase. 
     As such, when using a multi-carrier transmission method such as OFDM, and unlike single carrier transmission, symbols can be arranged in the frequency domain. Of course, the symbol arrangement method is not limited to those illustrated by  FIGS.  14 A and  14 B . Further examples are shown in  FIGS.  15 A,  15 B,  16 A, and  16 B . 
       FIGS.  15 A and  15 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  14 A and  14 B .  FIG.  15 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  15 B  illustrates a reordering method for the symbols of modulated signal z 2 .  FIGS.  15 A and  15 B  differ from  FIGS.  14 A and  14 B  in the reordering method applied to the symbols of modulated signal z 1  and the symbols of modulated signal z 2 . In  FIG.  15 B , symbols #0 through #5 are arranged at carriers  4  through  9 , symbols #6 though #9 are arranged at carriers  0  through  3 , and this arrangement is repeated for symbols #10 through #19. Here, as in  FIG.  14 B , symbol group  1502  shown in  FIG.  15 B  corresponds to one period (cycle) of symbols when the phase changing method of  FIG.  6    is used. 
       FIGS.  16 A and  16 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  14 A and  14 B .  FIG.  16 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  16 B  illustrates a reordering method for the symbols of modulated signal z 2 .  FIGS.  16 A and  16 B  differ from  FIGS.  14 A and  14 B  in that, while  FIGS.  14 A and  14 B  showed symbols arranged at sequential carriers,  FIGS.  16 A and  16 B  do not arrange the symbols at sequential carriers. Obviously, for  FIGS.  16 A and  16 B , different reordering methods may be applied to the symbols of modulated signal z 1  and to the symbols of modulated signal z 2  as in  FIGS.  15 A and  15 B . 
       FIGS.  17 A and  17 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from those of  FIGS.  14 A through  16 B .  FIG.  17 A  illustrates a reordering method for the symbols of modulated signal z 1  while  FIG.  17 B  illustrates a reordering method for the symbols of modulated signal z 2 . While  FIGS.  14 A through  16 B  show symbols arranged with respect to the frequency axis,  FIGS.  17 A and  17 B  use the frequency and time axes together in a single arrangement. 
     While  FIG.  6    describes an example where the change of phase is performed in a four slot period (cycle), the following example describes an eight slot period (cycle). In  FIGS.  17 A and  17 B , the symbol group  1702  is equivalent to one period (cycle) of symbols when the phase changing scheme is used (i.e., to eight symbols) such that symbol #0 is the symbol obtained by using the phase at timestamp u, symbol #1 is the symbol obtained by using the phase at timestamp u+1, symbol #2 is the symbol obtained by using the phase at timestamp u+2, symbol #3 is the symbol obtained by using the phase at timestamp u+3, symbol #4 is the symbol obtained by using the phase at timestamp u+4, symbol #5 is the symbol obtained by using the phase at timestamp u+5, symbol #6 is the symbol obtained by using the phase at timestamp u+6, and symbol #7 is the symbol obtained by using the phase at timestamp u+7. Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at timestamp u when x mod 8 equals 0, symbol #x is the symbol obtained by using the phase at timestamp u+1 when x mod 8 equals 1, symbol #x is the symbol obtained by using the phase at timestamp u+2 when x mod 8 equals 2, symbol #x is the symbol obtained by using the phase at timestamp u+3 when x mod 8 equals 3, symbol #x is the symbol obtained by using the phase at timestamp u+4 when x mod 8 equals 4, symbol #x is the symbol obtained by using the phase at timestamp u+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using the phase at timestamp u+6 when x mod 8 equals 6, and symbol #x is the symbol obtained by using the phase at timestamp u+7 when x mod 8 equals 7. In  FIGS.  17 A and  17 B  four slots along the time axis and two slots along the frequency axis are used for a total of 4×2=8 slots, in which one period (cycle) of symbols is arranged. Here, given m×n symbols per period (cycle) (i.e., m×n different phases are available for multiplication), then n slots (carriers) in the frequency domain and m slots in the time domain should be used to arrange the symbols of each period (cycle), such that m&gt;n. This is because the phase of direct waves fluctuates slowly in the time domain relative to the frequency domain. Accordingly, the present Embodiment performs a regular change of phase that reduces the influence of steady direct waves. Thus, the phase changing period (cycle) should preferably reduce direct wave fluctuations. Accordingly, m should be greater than n. Taking the above into consideration, using the time and frequency domains together for reordering, as shown in  FIGS.  17 A and  17 B , is preferable to using either of the frequency domain or the time domain alone due to the strong probability of the direct waves becoming regular. As a result, the effects of the present invention are more easily obtained. However, reordering in the frequency domain may lead to diversity gain due the fact that frequency-domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal. 
       FIGS.  18 A and  18 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  17 A and  17 B .  FIG.  18 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  18 B  illustrates a reordering method for the symbols of modulated signal z 2 . Much like  FIGS.  17 A and  17 B ,  FIGS.  18 A and  18 B  illustrate the use of the time and frequency axes, together. However, in contrast to  FIGS.  17 A and  17 B , where the frequency axis is prioritized and the time axis is used for secondary symbol arrangement,  FIGS.  18 A and  18 B  prioritize the rime axis and use the frequency axis for secondary symbol arrangement. In  FIG.  18 B , symbol group  1802  corresponds to one period (cycle) of symbols when the phase changing method is used. 
     In  FIGS.  17 A,  17 B,  18 A, and  18 B , the reordering method applied to the symbols of modulated signal z 1  and the symbols of modulated signal z 2  may be identical or may differ as like in  FIGS.  15 A and  15 B . Either approach allows good reception quality to be obtained. Also, in  FIGS.  17 A,  17 B,  18 A, and  18 B , the symbols may be arranged non-sequentially as in  FIGS.  16 A and  16 B . Either approach allows good reception quality to be obtained. 
       FIG.  22    indicates frequency on the horizontal axis and time on the vertical axis thereof, and illustrates an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from the above.  FIG.  22    illustrates a regular phase changing method using four slots, similar to timestamps u through u+3 from  FIG.  6   . The characteristic feature of  FIG.  22    is that, although the symbols are reordered with respect the frequency domain, when read along the time axis, a periodic shift of n (n=1 in the example of  FIG.  22   ) symbols is apparent. The frequency-domain symbol group  2210  in  FIG.  22    indicates four symbols to which the change of phase is applied at timestamps u through u+3 from  FIG.  6   . 
     Here, symbol #0 is obtained through a change of phase at timestamp u, symbol #1 is obtained through a change of phase at timestamp u+1, symbol #2 is obtained through a change of phase at timestamp u+2, and symbol #3 is obtained through a change of phase at timestamp u+3. 
     Similarly, for frequency-domain symbol group  2220 , symbol #4 is obtained through a change of phase at timestamp u, symbol #5 is obtained through a change of phase at timestamp u+1, symbol #6 is obtained through a change of phase at timestamp u+2, and symbol #7 is obtained through a change of phase at timestamp u+3. 
     The above-described change of phase is applied to the symbol at timestamp $ 1 . However, in order to apply periodic shifting with respect to the time domain, the following change of phases are applied to symbol groups  2201 ,  2202 ,  2203 , and  2204 . 
     For time-domain symbol group  2201 , symbol #0 is obtained through a change of phase at timestamp u, symbol #9 is obtained through a change of phase at timestamp u+1, symbol #18 is obtained through a change of phase at timestamp u+2, and symbol #27 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2202 , symbol #28 is obtained through a change of phase at timestamp u, symbol #1 is obtained through a change of phase at timestamp u+1, symbol #10 is obtained through a change of phase at timestamp u+2, and symbol #19 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2203 , symbol #20 is obtained through a change of phase at timestamp u, symbol #29 is obtained through a change of phase at timestamp u+1, symbol #2 is obtained through a change of phase at timestamp u+2, and symbol #11 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2204 , symbol #12 is obtained through a change of phase at timestamp u, symbol #21 is obtained through a change of phase at timestamp u+1, symbol #30 is obtained through a change of phase at timestamp u+2, and symbol #3 is obtained through a change of phase at timestamp u+3. 
     The characteristic feature of  FIG.  22    is seen in that, taking symbol #11 as an example, the two neighbouring symbols thereof having the same timestamp in the frequency domain (#10 and #12) are both symbols changed using a different phase than symbol #11, and the two neighbouring symbols thereof having the same carrier in the time domain (#2 and #20) are both symbols changed using a different phase than symbol #11. This holds not only for symbol #11, but also for any symbol having two neighbouring symbols in the frequency domain and the time domain. Accordingly, the change of phase is effectively carried out. This is highly likely to improve data reception quality as influence from regularizing direct waves is less prone to reception. 
     Although  FIG.  22    illustrates an example in which n=1, the invention is not limited in this manner. The same may be applied to a case in which n=3. Furthermore, although  FIG.  22    illustrates the realization of the above-described effects by arranging the symbols in the frequency domain and advancing in the time domain so as to achieve the characteristic effect of imparting a periodic shift to the symbol arrangement order, the symbols may also be randomly (or regularly) arranged to the same effect. 
     Embodiment 2 
     In Embodiment 1, described above, phase changing is applied to a weighted (precoded with a fixed precoding matrix) signal z(t). The following Embodiments describe various phase changing methods by which the effects of Embodiment 1 may be obtained. 
     In the above-described Embodiment, as shown in  FIGS.  3  and  6   , phase changer  317 B is configured to perform a change of phase on only one of the signals output by the weighting unit  600 . 
     However, phase changing may also be applied before precoding is performed by the weighting unit  600 . In addition to the components illustrated in FIG.  6 , the transmission device may also feature the weighting unit  600  before the phase changer  317 B, as shown in  FIG.  25   . 
     In such circumstances, the following configuration is possible. The phase changer  317 B performs a regular change of phase with respect to baseband signal s 2 ( t ), on which mapping has been performed according to a selected modulation scheme, and outputs s 2 ′( t )=s 2 ( t )y(t) (where y(t) varies over time t). The weighting unit  600  executes precoding on s 2 ′ t , outputs z 2 ( t )=W 2   s   2 ′( t ) (see Math. 42 (formula 42)) and the result is then transmitted. 
     Alternatively, phase changing may be performed on both modulated signals s 1 ( t ) and s 2 ( t ). As such, the transmission device is configured so as to include a phase changer taking both signals output by the weighting unit  600 , as shown in  FIG.  26   . 
     Like phase changer  317 B, phase changer  317 A performs regular a regular change of phase on the signal input thereto, and as such changes the phase of signal z 1 ′( t ) precoded by the weighting unit. Post-phase change signal z 1 ( t ) is then output to a transmitter. 
     However, the phase changing rate applied by the phase changers  317 A and  317 B varies simultaneously in order to perform the phase changing shown in  FIG.  26   . (The following describes a non-limiting example of the phase changing method.) For timestamp u, phase changer  317 A from  FIG.  26    performs the change of phase such that z 1 ( t )=y 1 ( t )z 1 ′( t ), while phase changer  317 B performs the change of phase such that z 2 ( t )=y 2 ( t )z 2 ′( t ). For example, as shown in  FIG.  26   , for timestamp u, y 1 (u)=e j0  and y 2 ( u )=e −jπ/2 , for timestamp u+1, y 1 (u+1)=e jπ/4  and y 2 ( u +1)=e −j3π/4 , and for timestamp u+k, y 1 (u+k)=e jkπ/4  and y 2 ( u +k)=e j(k3π/4−π/2) . Here, the regular phase changing period (cycle) may be the same for both phase changers  317 A and  317 B, or may vary for each. 
     Also, as described above, a change of phase may be performed before precoding is performed by the weighting unit. In such a case, the transmission device should be configured as illustrated in  FIG.  27    rather than as illustrated in  FIG.  26   . 
     When a change of phase is carried out on both modulated signals, each of the transmit signals is, for example, control information that includes information about the phase changing pattern. By obtaining the control information, the reception device knows the phase changing method by which the transmission device regularly varies the change, i.e., the phase changing pattern, and is thus able to demodulate (decode) the signals correctly. 
     Next, variants of the sample configurations shown in  FIGS.  6  and  25    are described with reference to  FIGS.  28  and  29   .  FIG.  28    differs from  FIG.  6    in the inclusion of phase change ON/OFF information  2800  and in that the change of phase is performed on only one of z 1 ′( t ) and z 2 ′( t ) (i.e., performed on one of z 1 ′( t ) and z 2 ′( t ), which have identical timestamps or a common frequency). Accordingly, in order to perform the change of phase on one of z 1 ′( t ) and z 2 ′( t ), the phase changers  317 A and  317 B shown in  FIG.  28    may each be ON, and performing the change of phase, or OFF, and not performing the change of phase. The phase change ON/OFF information  2800  is control information therefor. The phase change ON/OFF information  2800  is output by the signal processing method information generator  314  shown in  FIG.  3   . 
     Phase changer  317 A of  FIG.  28    changes the phase to produce z 1 ( t )=y 1 (t)z 1 ′( t ), while phase changer  317 B changes the phase to produce z 2 ( t )=y 2 ( t )z 2 ′( t ). 
     Here, a change of phase having a period (cycle) of four is, for example, applied to z 1 ′( t ). (Meanwhile, the phase of z 2 ′( t ) is not changed.) Accordingly, for timestamp u, y 1 (u)=e j0  and y 2 ( u )=1, for timestamp u+1, y 1 (u+1)=e jπ/2  and y 2 (u+1)=1, for timestamp u+2, y 1 (u+2)=e j/π  and y 2 (u+2)=1, and for timestamp u+3, y 1 (u+3)=e j3π/2  and y 2 (u+3)=1. 
     Next, a change of phase having a period (cycle) of four is, for example, applied to z 2 ′( t ). (Meanwhile, the phase of z 1 ′( t ) is not changed.) Accordingly, for timestamp u+4, y 1 (u+4)=1 and y 2 (u+4)=e j0 , for timestamp u+5, y 1 (u+5)=1 and y 2 (u+5)=e jπ/2 , for timestamp u+6, y 1 (u+6)=1 and y 2 (u+6)=e jπ , and for timestamp u+7, y 1 (u+7)=1 and y 2 ( u +7)=e j3π/2 . 
     Accordingly, given the above examples. 
     for any timestamp 8k, y 1 (8k)=e j0  and y 2 (8k)=1, 
     for any timestamp 8k+1, y 1 (8k+1)=e jπ/2  and y 2 (8k+1)=1, 
     for any timestamp 8k+2, y 1 (8k+2)=e jπ  and y 2 (8k+2)=1, 
     for any timestamp 8k+3, y 1 (8k+3)=e j3π/2  and y 2 (8k+3)=1, 
     for any timestamp 8k+4, y 1 (8k+4)=1 and y 2 (8k+4)=e j0 , 
     for any timestamp 8k+5, y 1 (8k+3)=1 and y 2 (8k+5)=e jπ/2 , 
     for any timestamp 8k+6, y 1 (8k+6)=1 and y 2 (8k+6)=e jπ , and 
     for any timestamp 8k+7, y 1 (8k+7)=1 and y 2 (8k+7)=e j3π/2 . 
     As described above, there are two intervals, one where the change of phase is performed on z 1 ′( t ) only, and one where the change of phase is performed on z 2 ′( t ) only. Furthermore, the two intervals form a phase changing period (cycle). While the above explanation describes the interval where the change of phase is performed on z 1 ′( t ) only and the interval where the change of phase is performed on z 2 ′( t ) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing a change of phase having a period (cycle) of four on z 1 ′( t ) only and then performing a change of phase having a period (cycle) of four on z 2 ′( t ) only, no limitation is intended in this manner. The changes of phase may be performed on z 1 ′( t ) and on z 2 ′( t ) in any order (e.g., the change of phase may alternate between being performed on z 1 ′( t ) and on z 2 ′( t ), or may be performed in random order). 
     Phase changer  317 A of  FIG.  29    changes the phase to produce s 1 =y 1 (t)s 1 ( t ), while phase changer  317 B changes the phase to produce s 2 ′( t )=y 2 (t)s 2 ( t ). 
     Here, a change of phase having a period (cycle) of four is, for example, applied to s 1 ( t ). (Meanwhile, s 2 ( t ) remains unchanged). Accordingly, for timestamp u, y 1 ( u )=e j0  and y 2 ( u )=1, for timestamp u+1, y 1 ( u +1)=e jπ/2  and y 2 ( u +1)=1, for timestamp u+2, y 1 ( u +2)=e jπ  and y 2 ( u +2)=1, and for timestamp u+3, y 1 ( u +3)=e j3π/2  and y 2 ( u +3)=1. 
     Next, a change of phase having a period (cycle) of four is, for example, applied to s 2 ( t ). (Meanwhile, s 1 ( t ) remains unchanged). Accordingly, for timestamp u+4, y 1 ( u +4)=1 and y 2 ( u +4)=e j0 , for timestamp u+5, y 1 ( u +5)=1 and y 2 ( u +5)=e jπ/2 , for timestamp u+6, y 1 ( u +6)=1 and y 2 ( u +6)=e jπ , and for timestamp u+7, y 1 ( u +7)=1 and y 2 ( u +7)=e j3π/2 . 
     Accordingly, given the above examples, 
     for any timestamp 8k, y 1 (8k)=e j0  and y 2 (8k)=1, 
     for any timestamp 8k+1, y 1 (8k+1)=e jπ/2  and y 2 (8k+1)=1, 
     for any timestamp 8k+2, y 1 (8k+2)=e jπ  and y 2 (8k+2)=1, 
     for any timestamp 8k+3, y 1 (8k+3)=e j3π/2  and y 2 (8k+3)=1, 
     for any timestamp 8k+4, y 1 (8k+4)=1 and y 2 (8k+4)=e j0 , 
     for any timestamp 8k+5, y 1 (8k+5)=1 and y 2 (8k+5)=e jπ/2 , 
     for any timestamp 8k+6, y 1 (8k+6)=1 and y 2 (8k+6)=e jπ , and 
     for any timestamp 8k+7, y 1 (8k+7)=1 and y 2 (8k+7)=e j3π/2 . 
     As described above, there are two intervals, one where the change of phase is performed on s 1 ( t ) only, and one where the change of phase is performed on s 2 ( t ) only. Furthermore, the two intervals form a phase changing period (cycle). Although the above explanation describes the interval where the change of phase is performed on s 1 ( t ) only and the interval where the change of phase is performed on s 2 ( t ) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing the change of phase having a period (cycle) of four on s 1 ( t ) only and then performing the change of phase having a period (cycle) of four on s 2 ( t ) only, no limitation is intended in this manner. The changes of phase may be performed on s 1 ( t ) and on s 2 ( t ) in any order (e.g., may alternate between being performed on s 1 ( t ) and on s 2 ( t ), or may be performed in random order). 
     Accordingly, the reception conditions under which the reception device receives each transmit signal z 1 ( t ) and z 2 ( t ) are equalized. By periodically switching the phase of the symbols in the received signals z 1 ( t ) and z 2 ( t ), the ability of the error corrected codes to correct errors may be improved, thus ameliorating received signal quality in the LOS environment. 
     Accordingly, Embodiment 2 as described above is able to produce the same results as the previously described Embodiment 1. 
     Although the present Embodiment used a single-carrier method, i.e., time domain phase changing, as an example, no limitation is intended in this regard. The same effects are also achievable using multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA (Single Carrier Frequency-Division Multiple Access), SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As previously described, while the present Embodiment explains the change of phase as changing the phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the phase changing method in the time domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing method of the present Embodiment is also applicable to a change of phase with respect to both the time domain and the frequency domain. 
     Accordingly, although  FIGS.  6 ,  25 ,  26 , and  27    illustrate changes of phase in the time domain, replacing time t with carrier f in each of  FIGS.  6 ,  25 ,  26 , and  27    corresponds to a change of phase in the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing the change of phase on time-frequency blocks. 
     Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. 
     Embodiment 3 
     Embodiments 1 and 2, described above, discuss regular changes of phase. Embodiment 3 describes a method of allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device. 
     Embodiment 3 concerns the symbol arrangement within signals obtained through a change of phase. 
       FIG.  31    illustrates an example of frame configuration for a portion of the symbols within a signal in the time-frequency domains, given a transmission method where a regular change of phase is performed for a multi-carrier method such as OFDM. 
     First, an example is explained in which the change of phase is performed one of two baseband signals, precoded as explained in Embodiment 1 (see  FIG.  6   ). 
     (Although  FIG.  6    illustrates a change of phase in the time domain, switching time t with carrier f in  FIG.  6    corresponds to a change of phase in the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing phase changes on time-frequency blocks.) 
       FIG.  31    illustrates the frame configuration of modulated signal z 2 ′, which is input to phase changer  317 B from  FIG.  12   . Each square represents one symbol (although both signals s 1  and s 2  are included for precoding purposes, depending on the precoding matrix, only one of signals s 1  and s 2  may be used). 
     Consider symbol  3100  at carrier  2  and timestamp $ 2  of  FIG.  31   . The carrier here described may alternatively be termed a sub-carrier. 
     Within carrier  2 , there is a very strong correlation between the channel conditions for symbol  3100 A at carrier  2 , timestamp $ 2  and the channel conditions for the time domain nearest-neighbour symbols to timestamp $ 2 , i.e., symbol  3013  at timestamp $ 1  and symbol  3101  at timestamp $ 3  within carrier  2 . 
     Similarly, for timestamp $ 2 , there is a very strong correlation between the channel conditions for symbol  3100  at carrier  2 , timestamp $ 2  and the channel conditions for the frequency-domain nearest-neighbour symbols to carrier  2 , i.e., symbol  3104  at carrier  1 , timestamp $ 2  and symbol  3104  at timestamp $ 2 , carrier  3 . 
     As described above, there is a very strong correlation between the channel conditions for symbol  3100  and the channel conditions for each symbol  3101 ,  3102 ,  3103 , and  3104 . 
     The present description considers N different phases (N being an integer, N≥2) for multiplication in a transmission method where the phase is regularly changed. The symbols illustrated in  FIG.  31    are indicated as e j0 , for example. This signifies that this symbol is signal z 2 ′ from  FIG.  6    having undergone a change in phase through multiplication by e j0 . That is, the values indicated in  FIG.  31    for each of the symbols are the values of y(t) from Math. 42 (formula 42), which are also the values of z 2 ( t )=y 2 ( t )z 2 ′( t ) described in Embodiment 2. 
     The present Embodiment takes advantage of the high correlation in channel conditions existing between neighbouring symbols in the frequency domain and/or neighbouring symbols in the time domain in a symbol arrangement enabling high data reception quality to be obtained by the reception device receiving the phase-changed symbols. 
     In order to achieve this high data reception quality, conditions #1 and #2 are necessary. 
     (Condition #1) 
     As shown in  FIG.  6   , for a transmission method involving a regular change of phase performed on precoded baseband signal z 2 ′ using multi-carrier transmission such as OFDM, time X, carrier Y is a symbol for transmitting data (hereinafter, data symbol), neighbouring symbols in the time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Y are also data symbols, and a different change of phase should be performed on precoded baseband signal z 2 ′ corresponding to each of these three data symbols, i.e., on precoded baseband signal z 2 ′ at time X, carrier Y, at time X−1, carrier Y and at time X+1, carrier Y. 
     (Condition #2) 
     As shown in  FIG.  6   , for a transmission method involving a regular change of phase performed on precoded baseband signal z 2 ′ using multi-carrier transmission such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the frequency domain, i.e., at time X, carrier Y−1 and at time X, carrier Y+1 are also data symbols, and a different change of phase should be performed on precoded baseband signal z 2 ′ corresponding to each of these three data symbols, i.e., on precoded baseband signal z 2 ′ at time X, carrier Y, at time X, carrier Y−1 and at time X, carrier Y+1. 
     Ideally, data symbols satisfying Condition #1 should be present. Similarly, data symbols satisfying Condition #2 should be present. 
     The reasons supporting Conditions #1 and #2 are as follows. 
     A very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above. 
     Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to phase relations despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding. 
     Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the frequency domain, as described above. 
     Accordingly, when three neighbouring symbols in the frequency domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding. 
     Combining Conditions #1 and #2, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition #3 can be derived. 
     (Condition #3) 
     As shown in  FIG.  6   , for a transmission method involving a regular change of phase performed on precoded baseband signal z 2 ′ using multi-carrier transmission such as OFDM, time X, carrier Y is a data symbol, neighbouring symbols in the time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Y are also data symbols, and neighbouring symbols in the frequency domain, i.e., at time X, carrier Y−1 and at time X, carrier Y+1 are also data symbols, and a different change in phase is performed on precoded baseband signal z 2 ′ corresponding to each of these five data symbols, i.e., on precoded baseband signal z 2 ′ at time X, carrier Y, at time X, carrier Y−1, at time X, carrier Y+1, at a time X−1, carrier Y, and at time X+1, carrier  Y.    
     Here, the different changes in phase are as follows. Phase changes are defined from 0 radians to 2π radians. For example, for time X, carrier Y, a phase change of e jθX,Y  is applied to precoded baseband signal z 2 ′ from  FIG.  6   , for time X−1, carrier Y, a phase change of e jθX−1,Y  is applied to precoded baseband signal z 2 ′ from  FIG.  6   , for time X+1, carrier Y, a phase change of e jθX+1,Y  is applied to precoded baseband signal z 2 ′ from  FIG.  6   , such that 0≤0 X,Y &lt;2π, 0≤θ X−1,Y &lt;2π, and 0≤θ X+1,Y &lt;2π, all units being in radians. Accordingly, for Condition #1, it follows that θ X,Y ≠θ X−1,Y , θ X,Y ≠θ X+1,Y , and that θ X,Y−1 ≠θ X,Y+1 . Similarly, for Condition #2, it follows that θ X,Y ≠θ X,Y−1 , θ X,Y ≠θ X,Y+1 , and that θ X,Y−1 ≠θ X,Y+1 . And, for Condition #3, it follows that θ X,Y ≠θ X−1,Y , θ X,Y ≠θ X+1,Y , θ X,Y ≠θ X,Y−1 , θ X,Y ≠θ X,Y−1 , θ X−1,Y ≠θ X+1,Y , θ X−1,Y ≠θ X,Y−1 , θ X−1,Y ≠θ X+1,Y , θ X+1,Y ≠θ X−1,Y , θ X+1,Y ≠θ X,Y+1 , and that θ X,Y−1 ≠θ X,Y+1 . 
     Ideally, data symbols satisfying Condition #3 should be present. 
       FIG.  31    illustrates an example of Condition #3 where symbol A corresponds to symbol  3100 . The symbols are arranged such that the phase by which precoded baseband signal z 2 ′ from  FIG.  6    is multiplied differs for symbol  3100 , for both neighbouring symbols thereof in the time domain  3101  and  3102 , and for both neighbouring symbols thereof in the frequency domain  3102  and  3104 . Accordingly, despite received signal quality degradation of symbol  3100  for the receiver, good signal quality is highly likely for the neighbouring signals, thus guaranteeing good signal quality after error correction. 
       FIG.  32    illustrates a symbol arrangement obtained through phase changes under these conditions. 
     As evident from  FIG.  32   , with respect to any data symbol, a different change in phase is applied to each neighbouring symbol in the time domain and in the frequency domain. As such, the ability of the reception device to correct errors may be improved. 
     In other words, in  FIG.  32   , when all neighbouring symbols in the time domain are data symbols, Condition #1 is satisfied for all Xs and all Ys. 
     Similarly, in  FIG.  32   , when all neighbouring symbols in the frequency domain are data symbols, Condition #2 is satisfied for all Xs and all Ys. 
     Similarly, in  FIG.  32   , when all neighbouring symbols in the frequency domain are data symbols and all neighbouring symbols in the time domain are data symbols, Condition #3 is satisfied for all Xs and all Ys. 
     The following describes an example in which a change of phase is performed on two precoded baseband signals, as explained in Embodiment 2 (see  FIG.  26   ). 
     When a change of phase is performed on precoded baseband signal z 1 ′ and precoded baseband signal z 2 ′ as shown in  FIG.  26   , several phase changing methods are possible. The details thereof are explained below. 
     Scheme 1 involves a change in phase of precoded baseband signal z 2 ′ as described above, to achieve the change in phase illustrated by  FIG.  32   . In  FIG.  32   , a change of phase having a period (cycle) of ten is applied to precoded baseband signal z 2 ′. However, as described above, in order to satisfy Conditions #1, #2, and #3, the change in phase applied to precoded baseband signal z 2 ′ at each (sub-)carrier varies over time. (Although such changes are applied in  FIG.  32    with a period (cycle) of ten, other phase changing methods are also possible.) Then, as shown in  FIG.  33   , the change in phase performed on precoded baseband signal z 1 ′ produces a constant value that is one-tenth of that of the change in phase performed on precoded baseband signal z 2 ′. In  FIG.  33   , for a period (cycle) (of change in phase performed on precoded baseband signal z 2 ′) including timestamp $ 1 , the value of the change in phase performed on precoded baseband signal z 1 ′ is e j0 . Then, for the next period (cycle) (of change in phase performed on precoded baseband signal z 2 ′) including timestamp $ 2 , the value of the change in phase performed on precoded baseband signal z 1 ′ is e jπ/9 , and so on. 
     The symbols illustrated in  FIG.  33    are indicated as e j0 , for example. This signifies that this symbol is signal z 1 ′ from  FIG.  26    to which a change in phase has been applied through multiplication by e j0 . That is, the values indicated in  FIG.  33    for each of the symbols are the values of z 1 ( t )=y 1 (t)z 1 ′(t) described in Embodiment 2 for y 1 (t). 
     As shown in  FIG.  33   , the change in phase performed on precoded baseband signal z 1 ′ produces a constant value that is one-tenth that of the change in phase performed on precoded baseband signal z 2 ′ such that the post-phase change value varies with the number of each period (cycle). (As described above, in  FIG.  33   , the value is e j0  for the first period (cycle), e jπ/9  for the second period (cycle), and so on.) 
     As described above, the change in phase performed on precoded baseband signal z 2 ′ has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the change in phase applied to precoded baseband signal z 1 ′ and to precoded baseband signal z 2 ′ into consideration. Accordingly, data reception quality may be improved for the reception device. 
     Scheme 2 involves a change in phase of precoded baseband signal z 2 ′ as described above, to achieve the change in phase illustrated by  FIG.  32   . In  FIG.  32   , a change of phase having a period (cycle) of ten is applied to precoded baseband signal z 2 ′. However, as described above, in order to satisfy Conditions #1, #2, and #3, the change in phase applied to precoded baseband signal z 2 ′ at each (sub-)carrier varies over time. (Although such changes are applied in  FIG.  32    with a period (cycle) of ten, other phase changing methods are also possible.) Then, as shown in  FIG.  30   , the change in phase performed on precoded baseband signal z 1 ′ differs from that performed on precoded baseband signal z 2 ′ in having a period (cycle) of three rather than ten. 
     The symbols illustrated in  FIG.  30    are indicated as e j0 , for example. This signifies that this symbol is signal z 1 ′ from  FIG.  26    to which a change in phase has been applied through multiplication by e j0 . That is, the values indicated in  FIG.  30    for each of the symbols are the values of z 1 ( t )=y 1 (t)z 1 ′( t ) described in Embodiment 2 for y 1 (t). 
     As described above, the change in phase performed on precoded baseband signal z 2 ′ has a period (cycle) of ten, but by taking the changes in phase applied to precoded baseband signal z 1 ′ and precoded baseband signal z 2 ′ into consideration, the period (cycle) can be effectively made equivalent to 30 for both precoded baseband signals z 1 ′ and z 2 ′. Accordingly, data reception quality may be improved for the reception device. An effective way of applying method 2 is to perform a change in phase on precoded baseband signal z 1 ′ with a period (cycle) of N and perform a change in phase on precoded baseband signal z 2 ′ with a period (cycle) of M such that N and M are coprime. As such, by taking both precoded baseband signals z 1 ′ and z 2 ′ into consideration, a period (cycle) of N×M is easily achievable, effectively making the period (cycle) greater when N and M are coprime. 
     The above describes an example of the phase changing method pertaining to Embodiment 3. The present invention is not limited in this manner. As explained for Embodiments 1 and 2, a change in phase may be performed with respect the frequency domain or the time domain, or on time-frequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases. 
     The same also applies to frames having a configuration other than that described above, where pilot symbols (SP symbols) and symbols transmitting control information are inserted among the data symbols. The details of the change in phase in such circumstances are as follows. 
       FIGS.  47 A and  47 B  illustrate the frame configuration of modulated signals (precoded baseband signals) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  47 A  illustrates the frame configuration of modulated signal (precoded baseband signal) z 1  or z 1 ′ while  FIG.  47 B  illustrates the frame configuration of modulated signal (precoded baseband signal) z 2 ′. In  FIGS.  47 A and  47 B,  4701    marks pilot symbols while  4702  marks data symbols. The data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed. 
       FIGS.  47 A and  47 B , like  FIG.  6   , indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal z 2 ′ (while no change of phase is performed on precoded baseband signal z 1 ). (Although  FIG.  6    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  6    corresponds to a change in phase with respect to the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  47 A and  47 B  for each of the symbols are the values of precoded baseband signal z 2 ′ after a change of phase is performed. No values are given for the symbols of precoded baseband signal z 1 ′ (z 1 ) as no change of phase is performed thereon. 
     The key point of  FIGS.  47 A and  47 B  is that a change of phase is performed on the data symbols of precoded baseband signal z 2 ′, i.e., on precoded symbols. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 2 ′. 
       FIGS.  48 A and  48 B  illustrate the frame configuration of modulated signals (precoded baseband signals) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain. FIG.  48 A illustrates the frame configuration of modulated signal (precoded baseband signal) z 1  or z 1 ′ while  FIG.  48 B  illustrates the frame configuration of modulated signal (precoded baseband signal) z 2 ′. In  FIGS.  48 A and  48 B,  4701    marks pilot symbols while  4702  marks data symbols. The data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed. 
       FIGS.  48 A and  48 B , like  FIG.  26   , indicate the arrangement of symbols when a change of phase is applied to precoded baseband signal z 1 ′ and to precoded baseband signal z 2 ′. (Although  FIG.  26    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  26    corresponds to a change in phase with respect to the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  48 A and  48 B  for each of the symbols are the values of precoded baseband signal z 1 ′ and z 2 ′ after a change of phase. 
     The key point of  FIGS.  48 A and  48 B  is that a change of phase is performed on the data symbols of precoded baseband signal z 1 ′, that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal z 2 ′, that is, on the precoded symbols thereof. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 1 ′, nor on the pilot symbols inserted in z 2 ′. 
       FIGS.  49 A and  49 B  illustrate the frame configuration of modulated signals (precoded baseband signals) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  49 A  illustrates the frame configuration of modulated signal (precoded baseband signal) z 1  or z 1 ′ while  FIG.  49 B  illustrates the frame configuration of modulated signal (precoded baseband signal) z 2 ′. In  FIGS.  49 A and  49 B,  4701    marks pilot symbols,  4702  marks data symbols, and  4901  marks null symbols for which the in-phase component of the baseband signal I=0 and the quadrature component Q=0. As such, data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed.  FIGS.  49 A and  49 B  differ from  FIGS.  47 A and  47 B  in the configuration method for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal z 1 ′ are null symbols in modulated signal z 2 ′. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal z 2 ′ are null symbols in modulated signal z 1 ′. 
       FIGS.  49 A and  49 B , like  FIG.  6   , indicate the arrangement of symbols when a change in phase is applied to precoded baseband signal z 2 ′ (while no change of phase is performed on precoded baseband signal z 1 ). (Although  FIG.  6    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  6    corresponds to a change in phase with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  49 A and  49 B  for each of the symbols are the values of precoded baseband signal z 2 ′ after a change of phase is performed. No values are given for the symbols of precoded baseband signal z 1 ′ (z 1 ) as no change of phase is performed thereon. 
     The key point of  FIGS.  49 A and  49 B  is that a change of phase is performed on the data symbols of precoded baseband signal z 2 ′, i.e., on precoded symbols. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 2 ′. 
       FIGS.  50 A and  50 B  illustrate the frame configuration of modulated signals (precoded baseband signals) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  50 A  illustrates the frame configuration of modulated signal (precoded baseband signal) z 1  or z 1 ′ while  FIG.  50 B  illustrates the frame configuration of modulated signal (precoded baseband signal) z 2 ′. In  FIGS.  50 A and  50 B,  4701    marks pilot symbols,  4702  marks data symbols, and  4901  marks null symbols for which the in-phase component of the baseband signal I=0 and the quadrature component Q=0. As such, data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed.  FIGS.  50 A and  50 B  differ from  FIGS.  48 A and  48 B  in the configuration method for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal z 1 ′ are null symbols in modulated signal z 2 ′. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal z 2 ′ are null symbols in modulated signal z 1 ′. 
       FIGS.  50 A and  50 B , like  FIG.  26   , indicate the arrangement of symbols when a change of phase is applied to precoded baseband signal z 1 ′ and to precoded baseband signal z 2 ′. (Although  FIG.  26    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  26    corresponds to a change in phase with respect to the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  50 A and  50 B  for each of the symbols are the values of precoded baseband signal z 1 ′ and z 2 ′ after the change in phase. 
     The key point of  FIGS.  50 A and  50 B  is that a change of phase is performed on the data symbols of precoded baseband signal z 1 ′, that is, on the precoded symbols thereof, and on the data symbols of precoded baseband signal z 2 ′, that is, on the precoded symbols thereof. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 1 ′, nor on the pilot symbols inserted in z 2 ′. 
       FIG.  51    illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the frame configuration of  FIGS.  47 A,  47 B,  49 A, and  49 B . Components thereof performing the same operations as those of  FIG.  4    use the same reference symbols thereas. 
     In  FIG.  51   , the weighting units  308 A and  308 B and phase changer  317 B only operate at times indicated by the frame configuration signal  313  as corresponding to data symbols. 
     In  FIG.  51   , a pilot symbol generator  5101  (that also generates null symbols) outputs baseband signals  5102 A and  5102 B for a pilot symbol whenever the frame configuration signal  313  indicates a pilot symbol (and a null symbol). 
     Although not indicated in the frame configurations from  FIGS.  47 A through  50 B , when precoding (or phase rotation) is not performed, such as when transmitting a modulated signal using only one antenna (such that the other antenna transmits no signal) or when using a space-time coding transmission method (particularly, space-time block coding) to transmit control information symbols, then the frame configuration signal  313  takes control information symbols  5104  and control information  5103  as input. When the frame configuration signal  313  indicates a control information symbol, baseband signals  5102 A and  5102 B thereof are output. 
     Wireless units  310 A and  310 B of  FIG.  51    take a plurality of baseband signals as input and select a desired baseband signal according to the frame configuration signal  313 . The wireless units  310 A and  310 B then apply OFDM signal processing and output modulated signals  311 A and  311 B conforming to the frame configuration. 
       FIG.  52    illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the frame configuration of  FIGS.  48 A,  48 B,  50 A, and  50 B . Components thereof performing the same operations as those of  FIGS.  4  and  51    use the same reference symbols thereas.  FIG.  51    features an additional phase changer  317 A that only operates when the frame configuration signal  313  indicates a data symbol. At all other times, the operations are identical to those explained for  FIG.  51   . 
       FIG.  53    illustrates a sample configuration of a transmission device that differs from that of  FIG.  51   . The following describes the points of difference. As shown in  FIG.  53   , phase changer  317 B takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  317 B performs the change in phase on precoded baseband signal  316 B. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  317 B pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     A selector  5301  takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal  313  for output. 
       FIG.  54    illustrates a sample configuration of a transmission device that differs from that of  FIG.  52   . The following describes the points of difference. As shown in  FIG.  54   , phase changer  317 B takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  317 B performs the change in phase on precoded baseband signal  316 B. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  317 B pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     Similarly, as shown in  FIG.  54   , phase changer  5201  takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  5201  performs the change in phase on precoded baseband signal  309 A. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  5201  pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     The above explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in this manner. When symbols are transmitted using methods other than precoding, such as single-antenna transmission or transmission using space-time block coding, not performing a change of phase is important. Conversely, performing a change of phase on symbols that have been precoded is the key point of the present invention. 
     Accordingly, a characteristic feature of the present invention is that the change of phase is not performed on all symbols within the frame configuration in the time-frequency domain, but only performed on signals that have been precoded. 
     Embodiment 4 
     Embodiments 1 and 2, described above, discuss a regular change of phase. Embodiment 3, however, discloses performing a different change of phase on neighbouring symbols. 
     The present Embodiment describes a phase changing method that varies according to the modulation scheme and the coding rate of the error-correcting codes used by the transmission device. 
     Table 1, below, is a list of phase changing method settings corresponding to the settings and parameters of the transmission device. 
     
       
         
           
               
               
               
               
             
               
                 TABLE 1 
               
               
                   
               
               
                 No. of  
                   
                   
                   
               
               
                 Modulated 
                   
                   
                 Phase 
               
               
                 Transmission 
                 Modulation  
                   
                 Changing 
               
               
                 Signals 
                 Scheme 
                 Coding Rate 
                 Pattern 
               
               
                   
               
             
            
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: 1/2, #2 2/3 
                 #1: —, #2: A 
               
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: 1/2, #2: 3/4 
                 #1: A, #2: B 
               
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: 2/3, #2: 3/5 
                 #1: A, #2: C 
               
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: 2/3, #2: 2/3 
                 #1: C, #2: — 
               
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: 3/3, #2: 5/6 
                 #1: D, #2: E 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: 1/2, #2: 2/3 
                 #1: B, #2: A 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: 1/2, #2: 3/4 
                 #1: A, #2: C 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: 1/2, #2: 3/5 
                 #1: —, #2: E 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: 2/3, #2: 3/4 
                 #1: D, #2: — 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: 2/3, #2: 5/6 
                 #1: D, #2: B 
               
               
                 2 
                 #1: 16-QAM, #2: 
                 #1: 1/2, #2: 2/3 
                 #1: —, #2: E 
               
               
                   
                 16-QAM 
                   
                   
               
               
                 . 
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
                 . 
               
               
                   
               
            
           
         
       
     
     In Table 1, #1 denotes modulated signal s 1  from Embodiment 1 described above (baseband signal s 1  modulated with the modulation scheme set by the transmission device) and #2 denotes modulated signal s 2  (baseband signal s 2  modulated with the modulation scheme set by the transmission device). The coding rate column of Table 1 indicates the coding rate of the error-correcting codes for modulation schemes #1 and #2. The phase changing pattern column of Table 1 indicates the phase changing method applied to precoded baseband signals z 1  (z 1 ′) and z 2  (z 2 ′), as explained in Embodiments 1 through 3. Although the phase changing patterns are labelled A, B, C, D, E, and so on, this refers to the phase change degree applied, for example, in a phase changing pattern given by Math. 46 (formula 46) and Math. 47 (formula 47), above. In the phase changing pattern column of Table 1, the dash signifies that no change of phase is applied. 
     The combinations of modulation scheme and coding rate listed in Table 1 are examples. Other modulation schemes (such as 128-QAM and 256-QAM) and coding rates (such as ⅞) not listed in Table 1 may also be included. Also, as described in Embodiment 1, the error-correcting codes used for s 1  and s 2  may differ (Table 1 is given for cases where a single type of error-correcting codes is used, as in  FIG.  4   ). Furthermore, the same modulation scheme and coding rate may be used with different phase changing patterns. The transmission device transmits information indicating the phase changing patterns to the reception device. The reception device specifies the phase changing pattern by cross-referencing the information and Table 1, then performs demodulation and decoding. When the modulation scheme and error-correction method determine a unique phase changing pattern, then as long as the transmission device transmits the modulation scheme and information regarding the error-correction method, the reception device knows the phase changing pattern by obtaining that information. As such, information pertaining to the phase changing pattern is not strictly necessary. 
     In Embodiments 1 through 3, the change of phase is applied to precoded baseband signals. However, the amplitude may also be modified along with the phase in order to apply periodical, regular changes. Accordingly, an amplification modification pattern regularly modifying the amplitude of the modulated signals may also be made to conform to Table 1. In such circumstances, the transmission device should include an amplification modifier that modifies the amplification after weighting unit  308 A or weighting unit  308 B from  FIG.  3  or  4   . In addition, amplification modification may be performed on only one of or on both of the precoded baseband signals z 1 ( t ) and z 2 ( t ) (in the former case, the amplification modifier is only needed after one of weighting unit  308 A and  308 B). 
     Furthermore, although not indicated in Table 1 above, the mapping scheme may also be regularly modified by the mapper, without a regular change of phase. 
     That is, when the mapping method for modulated signal s 1 ( t ) is 16-QAM and the mapping method for modulated signal s 2 ( t ) is also 16-QAM, the mapping method applied to modulated signal s 2 ( t ) may be regularly changed as follows: from 16-QAM to 16-APSK, to 16-QAM in the I-Q plane, to a first mapping method producing signal point distribution unlike 16-APSK, to 16-QAM in the I-Q plane, to a second mapping method producing signal point distribution unlike 16-APSK, and so on. As such, the data reception quality can be improved for the reception device, much like the results obtained by a regular change of phase described above. 
     In addition, the present invention may use any combination of methods for a regular change of phase, mapping method, and amplitude, and the transmit signal may transmit with all of these taken into consideration. 
     The present Embodiment may be realized using single-carrier methods as well as multi-carrier methods. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDM, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As described above, the present Embodiment describes changing the phase, amplitude, and mapping methods by performing phase, amplitude, and mapping method modifications with respect to the time domain t. However, much like Embodiment 1, the same changes may be carried out with respect to the frequency domain. That is, considering the phase, amplitude, and mapping method modification in the time domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) frequency) leads to phase, amplitude, and mapping method modification applicable to the frequency domain. Also, the phase, amplitude, and mapping method modification of the present Embodiment is also applicable to phase, amplitude, and mapping method modification in both the time domain and the frequency domain. 
     Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner. 
     Embodiment A1 
     The present Embodiment describes a method of regularly changing the phase when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC (Quasi-Cyclic) LDPC Codes (not only QC-LDPC but also LDPC codes may be used), concatenated LDPC and BCH (Bose-Chaudhuri-Hocquenghem) codes, Turbo codes or Duo-Binary Turbo codes using tail-biting, and so on. The following example considers a case where two streams s 1  and s 2  are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. 
       FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used.  FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  4   , and the transmission device has only one encoder. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  34   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     Then, given that the transmission device from  FIG.  4    transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s 1  and the other 1500 symbols are assigned to s 2 . As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s 1  and s 2 . 
     By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up each coded block, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up each coded block. 
     The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changer of the transmission device from  FIG.  4    (equivalent to the period (cycle) from Embodiments 1 through 4) (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase with a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). These five phase changing values (or phase changing sets) are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ], PHASE[ 3 ], and PHASE[ 4 ]. 
     For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[ 0 ] is used on 300 slots, PHASE[ 1 ] is used on 300 slots, PHASE[ 2 ] is used on 300 slots, PHASE[ 3 ] is used on 300 slots, and PHASE[ 4 ] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. 
     Further still, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, PHASE[ 0 ] is used on 150 slots, PHASE[ 1 ] is used on 150 slots, PHASE[ 2 ] is used on 150 slots, PHASE[ 3 ] is used on 150 slots, and PHASE[ 4 ] is used on 150 slots. 
     Further still, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, PHASE[ 0 ] is used on 100 slots, PHASE[ 1 ] is used on 100 slots, PHASE[ 2 ] is used on 100 slots, PHASE[ 3 ] is used on 100 slots, and PHASE[ 4 ] is used on 100 slots. 
     As described above, a method for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bits making up a single coded block, PHASE[ 0 ] is used on K 0  slots, PHASE[ 1 ] is used on K 1  slots, PHASE[i] is used on K i  slots (where i=0, 1, 2 . . . N−1), and PHASE[N−1] is used on K N−1  slots, such that Condition #A01 is met. 
     (Condition #A01) 
     K 0 =K 1  . . . =K i =K N−1 . That is, K a =K b (∀a and ∀b where a, b,=0, 1, 2 . . . N−1; (a and b being integers between 0 and N−1), a≠b). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition #A01 is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #A01 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #A01. 
     (Condition #A02) 
     The difference between K a  and K b  is 0 or 1. That is, |K a −K b | is 0 or 1(∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b) 
       FIG.  35    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used.  FIG.  35    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  3    and  FIG.  12   , and the transmission device has two encoders. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  35   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     The transmission device from  FIG.  3    and the transmission device from  FIG.  12    each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. Accordingly, when the modulation scheme is QPSK, two coded blocks drawn from s 1  and s 2  are transmitted within the same interval, e.g., a first coded block drawn from s 1  is transmitted, then a second coded block drawn from s 2  is transmitted. As such, 3000 slots are needed in order to transmit the first and second coded blocks. 
     By the same reasoning, when the modulation scheme is 16-QAM, 1500 slots are needed to transmit all of the bits making up the two coded blocks, and when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks 
     The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changer of the transmission device from  FIGS.  3  and  12    (equivalent to the period (cycle) from Embodiments 1 through 4) (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase with a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). These five phase changing values (or phase changing sets) are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ], PHASE[ 3 ], and PHASE[ 4 ]. 
     For the above-described 3000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is QPSK, PHASE[ 0 ] is used on 600 slots, PHASE[ 1 ] is used on 600 slots, PHASE[ 2 ] is used on 600 slots, PHASE[ 3 ] is used on 600 slots, and PHASE[ 4 ] is used on 600 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 600 times, PHASE[ 1 ] is used on slots 600 times, PHASE[ 2 ] is used on slots 600 times, PHASE[ 3 ] is used on slots 600 times, and PHASE[ 4 ] is used on slots 600 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 600 times, PHASE[ 1 ] is used on slots 600 times, PHASE[ 2 ] is used on slots 600 times, PHASE[ 3 ] is used on slots 600 times, and PHASE[ 4 ] is used on slots 600 times. 
     Similarly, for the above-described 1500 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 16-QAM, PHASE[ 0 ] is used on 300 slots, PHASE[ 1 ] is used on 300 slots, PHASE[ 2 ] is used on 300 slots, PHASE[ 3 ] is used on 300 slots, and PHASE[ 4 ] is used on 300 slots. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 300 times, PHASE[ 1 ] is used on slots 300 times, PHASE[ 2 ] is used on slots 300 times, PHASE[ 3 ] is used on slots 300 times, and PHASE[ 4 ] is used on slots 300 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 300 times, PHASE[ 1 ] is used on slots 300 times, PHASE[ 2 ] is used on slots 300 times, PHASE[ 3 ] is used on slots 300 times, and PHASE[ 4 ] is used on slots 300 times. 
     Similarly, for the above-described 1000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 64-QAM, PHASE[ 0 ] is used on 200 slots, PHASE[ 1 ] is used on 200 slots, PHASE[ 2 ] is used on 200 slots, PHASE[ 3 ] is used on 200 slots, and PHASE[ 4 ] is used on 200 slots. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 200 times, PHASE[ 1 ] is used on slots 200 times, PHASE[ 2 ] is used on slots 200 times, PHASE[ 3 ] is used on slots 200 times, and PHASE[ 4 ] is used on slots 200 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 200 times, PHASE[ 1 ] is used on slots 200 times, PHASE[ 2 ] is used on slots 200 times, PHASE[ 3 ] is used on slots 200 times, and PHASE[ 4 ] is used on slots 200 times. 
     As described above, a method for regularly changing the phase requires the preparation of phase changing values (or phase changing sets) expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], PHASE[N−1]. As such, in order to transmit all of the bits making up two coded blocks, PHASE[ 0 ] is used on K 0  slots, PHASE[ 1 ] is used on K 1  slots, PHASE[i] is used on K i  slots (where i=0, 1, 2 . . . N−1), and PHASE[N−1] is used on K N−1  slots, such that Condition #A03 is met. 
     (Condition #A03) 
     K 0 =K 1  . . . =K i = . . . K N−1 . That is, K a =K b (∀a and ∀b where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     Further, in order to transmit all of the bits making up the first coded block, PHASE[ 0 ] is used K 0,1  times, PHASE[ 1 ] is used K 1,1  times, PHASE[i] is used K i,1  times (where i=0, 1, 2 . . . N−1), and PHASE[N−1] is used K N−1,1  times, such that Condition #A04 is met.
 
(Condition #A04)
 
K 0,1 =K 1,1 = . . . K N−1,1 . That is, K a,1 =K b,1 (∀a and ∀b where a, b,=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1),a≠b).
 
     Furthermore, in order to transmit all of the bits making up the second coded block, PHASE[ 0 ] is used K 0,2  times, PHASE[ 1 ] is used K 1,2  times, PHASE[i] is used K 1,2  times (where i=0, 1, 2 . . . N−1), and PHASE[N−1] is used K N−1,2  times, such that Condition #A05 is met. 
     (Condition #A05) 
     K 0,2 =K 1,2 = . . . K 1,2 = . . . K N−1,2 . That is, K a,2 =K b,2  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition #A03, #A04, and #A05 is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbol (though some may happen to use the same number), Conditions #A03, #A04, and #A05 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #A03, #A04, and #A05. 
     (Condition #A06) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b) 
     (Condition #A07) 
     The difference between K a,1  and K b,1  satisfies 0 or 1. That is, |K a,1 −K b,1 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b) 
     (Condition #A08) 
     The difference between K a,2  and K b,2  satisfies 0 or 1. That is, |K a,2 −K b,2 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b) 
     As described above, bias among the phases being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase of multiplication. As such, data reception quality may be improved for the reception device. 
     In the present Embodiment, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the method for a regular change of phase. As such, N phase changing values (or phase changing sets) PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], and PHASE[N−1] are prepared. However, schemes exist for reordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) may also change the phases of blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing method with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the above-described conditions are satisfied, great quality data reception improvements are realizable for the reception device. 
     Furthermore, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, space-time block coding schemes, single-stream transmission, and schemes using a regular change of phase (the transmission schemes described in Embodiments 1 through 4), the transmission device (broadcaster, base station) may select any one of these transmission schemes. 
     As described in Non-Patent Literature 3, spatial multiplexing MIMO methods involve transmitting signals s 1  and s 2 , which are mapped using a selected modulation scheme, on each of two different antennas. As described in Embodiments 1 through 4, MIMO methods using a fixed precoding matrix involve performing precoding only (with no change of phase). Further, space-time block coding methods are described in Non-Patent Literature 9, 16, and 17. Single-stream transmission methods involve transmitting signal s 1 , mapped with a selected modulation scheme, from an antenna after performing predetermined processing. 
     Schemes using multi-carrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multi-carrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, space-time block coding schemes, single-stream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub-)carrier group are preferably used to realize the present Embodiment. 
     When a change of phase is performed, then for example, a phase changing value for PHASE[i] of X radians is performed on only one precoded baseband signal, the phase changers of  FIGS.  3 ,  4 ,  5 ,  12 ,  25 ,  29 ,  51 , and  53    multiplies precoded baseband signal z 2 ′ by e jX . Then, when a change of phase by, for example, a phase changing set for PHASE[i] of X radians and Y radians is performed on both precoded baseband signals, the phase changers from  FIGS.  26 ,  27 ,  28 ,  52 , and  54    multiply precoded baseband signal z 2 ′ by e jX  and multiply precoded baseband signal z 1 ′ by e jY . 
     Embodiment B1 
     The following describes a sample configuration of an application of the transmission methods and reception methods discussed in the above embodiments and a system using the application. 
       FIG.  36    illustrates the configuration of a system that includes devices executing transmission methods and reception methods described in the above Embodiments. As shown in  FIG.  36   , the devices executing transmission methods and reception methods described in the above Embodiments include various receivers such as a broadcaster, a television  3611 , a DVD recorder  3612 , a STB (set-top box)  3613 , a computer  3620 , a vehicle-mounted television  3641 , a mobile phone  3630  and so on within a digital broadcasting system  3600 . Specifically, the broadcaster  3601  uses a transmission method discussed in the above-described Embodiments to transmit multiplexed data, in which video, audio, and other data are multiplexed, over a predetermined transmission band. 
     The signals transmitted by the broadcaster  3601  are received by an antenna (such as antenna  3660  or  3640 ) embedded within or externally connected to each of the receivers. Each receiver obtains the multiplexed data by using reception methods discussed in the above-described Embodiments to demodulate the signals received by the antenna. Accordingly, the digital broadcasting system  3600  is able to realize the effects of the present invention, as discussed in the above-described Embodiments. 
     The video data included in the multiplexed data are coded with a video coding method compliant with a standard such as MPEG-2 (Moving Picture Experts Group), MPEG4-AVC (Advanced Video Coding), VC-1, or the like. The audio data included in the multiplexed data are encoded with an audio coding method compliant with a standard such as Dolby AC-3 (Audio Coding), Dolby Digital Plus, MLP (Meridian Lossless Packing), DTS (Digital Theatre Systems), DTS-HD, Linear PCM (Pulse-Code Modulation), or the like. 
       FIG.  37    illustrates the configuration of a receiver  7900  that executes a reception method described in the above-described Embodiments. The receiver  3700  corresponds to a receiver included in one of the television  3611 , the DVD recorder  3612 , the STB  3613 , the computer  3620 , the vehicle-mounted television  3641 , the mobile phone  3630  and so on from  FIG.  36   . The receiver  3700  includes a tuner  3701  converting a high-frequency signal received by an antenna  3760  into a baseband signal, and a demodulator  3702  demodulating the baseband signal so converted to obtain the multiplexed data. The demodulator  3702  executes a reception method discussed in the above-described Embodiments, and thus achieves the effects of the present invention as explained above. 
     The receiver  3700  further includes a stream interface  3720  that demultiplexes the audio and video data in the multiplexed data obtained by the demodulator  3702 , a signal processor  3704  that decodes the video data obtained from the demultiplexed video data into a video signal by applying a video decoding method corresponding thereto and decodes the audio data obtained from the demultiplexed audio data into an audio signal by applying an audio decoding method corresponding thereto, an audio output unit  3706  that outputs the decoded audio signal through a speaker or the like, and a video display unit  3707  that outputs the decoded video signal on a display or the like. 
     When, for example, a user uses a remote control  3750 , information for a selected channel (selected (television) program or audio broadcast) is transmitted to an operation input unit  3710 . Then, the receiver  3700  performs processing on the received signal received by the antenna  3760  that includes demodulating the signal corresponding to the selected channel, performing error-correcting decoding, and so on, in order to obtain the received data. At this point, the receiver  3700  obtains control symbol information that includes information on the transmission method (the transmission method, modulation scheme, error-correction method, and so on from the above-described Embodiments) (as described using  FIGS.  5  and  41   ) from control symbols included the signal corresponding to the selected channel. As such, the receiver  3700  is able to correctly set the reception operations, demodulation scheme, error-correction method and so on, thus enabling the data included in the data symbols transmitted by the broadcaster (base station) to be obtained. Although the above description is given for an example of the user using the remote control  3750 , the same operations apply when the user presses a selection key embedded in the receiver  3700  to select a channel. 
     According to this configuration, the user is able to view programs received by the receiver  3700 . 
     The receiver  3700  pertaining to the present Embodiment further includes a drive  3708  that may be a magnetic disk, an optical disc, a non-volatile semiconductor memory, or a similar recording medium. The receiver  3700  stores data included in the demultiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding (in some circumstances, the data obtained through demodulation by the demodulator  3702  may not be subject to error correction. Also, the receiver  3700  may perform further processing after error correction. The same hereinafter applies to similar statements concerning other components), data corresponding to such data (e.g., data obtained through compression of such data), data obtained through audio and video processing, and so on, on the drive  3708 . Here, an optical disc is a recording medium, such as DVD (Digital Versatile Disc) or BD (Blu-ray Disc), that is readable and writable with the use of a laser beam. A magnetic disk is a floppy disk, a hard disk, or similar recording medium on which information is storable through the use of magnetic flux to magnetize a magnetic body. A non-volatile semiconductor memory is a recording medium, such as flash memory or ferroelectric random access memory, composed of semiconductor element(s). Specific examples of non-volatile semiconductor memory include an SD card using flash memory and a Flash SSD (Solid State Drive). Naturally, the specific types of recording media mentioned herein are merely examples. Other types of recording mediums may also be used. 
     According to this structure, the user is able to record and store programs received by the receiver  3700 , and is thereby able to view programs at any given time after broadcasting by reading out the recorded data thereof. 
     Although the above explanations describe the receiver  3700  storing multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding on the drive  3708 , a portion of the data included in the multiplexed data may instead be extracted and recorded. For example, when data broadcasting services or similar content is included along with the audio and video data in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, the audio and video data may be extracted from the multiplexed data demodulated by the demodulator  3702  and stored as new multiplexed data. Furthermore, the drive  3708  may store either the audio data or the video data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding as new multiplexed data. The aforementioned data broadcasting service content included in the multiplexed data may also be stored on the drive  3708 . 
     Furthermore, when a television, recording device (e.g., a DVD recorder, BD recorder HDD recorder, SD card, or similar), or mobile phone incorporating the receiver  3700  of the present invention receives multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding that includes data for correcting bugs in software used to operate the television or recording device, for correcting bugs in software for preventing personal information and recorded data from being leaked, and so on, such software bugs may be corrected by installing the data on the television or recording device. As such, bugs in the receiver  3700  are corrected through the inclusion of data for correcting bugs in the software of the receiver  3700 . Accordingly, the television, recording device, or mobile phone incorporating the receiver  3700  may be made to operate more reliably. 
     Here, the process of extracting a portion of the data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding is performed by, for example, the stream interface  3703 . Specifically, the stream interface  3703 , demultiplexes the various data included in the multiplexed data demodulated by the demodulator  3702 , such as audio data, video data, data broadcasting service content, and so on, as instructed by a non-diagrammed controller such as a CPU. The stream interface  3703  then extracts and multiplexes only the indicated demultiplexed data, thus generating new multiplexed data. The data to be extracted from the demultiplexed data may be determined by the user or may be determined in advance according to the type of recording medium. 
     According to such a structure, the receiver  3700  is able to extract and record only the data needed in order to view the recorded program. As such, the amount of data to be recorded can be reduced. 
     Although the above explanation describes the drive  3708  as storing multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, the video data included in the multiplexed data so obtained may be converted by using a different video coding method than the original video coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The drive  3708  may then store the converted video data as new multiplexed data. Here, the video coding method used to generate the new video data may conform to a different standard than that used to generate the original video data. Alternatively, the same video coding method may be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding may be converted by using a different audio coding method than the original audio coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The drive  3708  may then store the converted audio data as new multiplexed data. 
     Here, the process by which the audio or video data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding is converted so as to reduce the amount of data or the bit rate thereof is performed by, for example, the stream interface  3703  or the signal processor  3704 . Specifically, the stream interface  3703  demultiplexes the various data included in the multiplexed data demodulated by the demodulator  3702 , such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller such as a CPU. The signal processor  3704  then performs processing to convert the video data so demultiplexed by using a different video coding method than the original video coding method applied thereto, and performs processing to convert the audio data so demultiplexed by using a different video coding method than the original audio coding method applied thereto. As instructed by the controller, the stream interface  3703  then multiplexes the converted audio and video data, thus generating new multiplexed data. The signal processor  3704  may, in accordance with instructions from the controller, performing conversion processing on either the video data or the audio data, alone, or may perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion may be specified by the user or determined in advance according to the type of recording medium. 
     According to such a structure, the receiver  3700  is able to modify the amount of data or the bitrate of the audio and video data for storage according to the data storage capacity of the recording medium, or according to the data reading or writing speed of the drive  3708 . Therefore, programs can be stored on the drive despite the storage capacity of the recording medium being less than the amount of multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, or the data reading or writing speed of the drive being lower than the bit rate of the demultiplexed data obtained through demodulation by the demodulator  3702 . As such, the user is able to view programs at any given time after broadcasting by reading out the recorded data. 
     The receiver  3700  further includes a stream output interface  3709  that transmits the multiplexed data demultiplexed by the demodulator  3702  to external devices through a communications medium  3730 . The stream output interface  3709  may be, for example, a wireless communication device transmitting modulated multiplexed data to an external device using a wireless transmission method conforming to a wireless communication standard such as Wi-Fi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGiG, WirelessHD, Bluetooth™, ZigBee™, and so on through a wireless medium (corresponding to the communications medium  3730 ). The stream output interface  3709  may also be a wired communication device transmitting modulated multiplexed data to an external device using a communication method conforming to a wired communication standard such as Ethernet™, USB (Universal Serial Bus), PLC (Power Line Communication), HDMI (High-Definition Multimedia Interface) and so on through a wired transmission path (corresponding to the communications medium  3730 ) connected to the stream output interface  3709 . 
     According to this configuration, the user is able to use an external device with the multiplexed data received by the receiver  3700  using the reception method described in the above-described Embodiments. The usage of multiplexed data by the user here includes use of the multiplexed data for real-time viewing on an external device, recording of the multiplexed data by a recording unit included in an external device, and transmission of the multiplexed data from an external device to a yet another external device. 
     Although the above explanations describe the receiver  3700  outputting multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding through the stream output interface  3709 , a portion of the data included in the multiplexed data may instead be extracted and output. For example, when data broadcasting services or similar content is included along with the audio and video data in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, the audio and video data may be extracted from the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, multiplexed and output by the stream output interface  3709  as new multiplexed data. In addition, the stream output interface  3709  may store either the audio data or the video data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding as new multiplexed data. 
     Here, the process of extracting a portion of the data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding is performed by, for example, the stream interface  3703 . Specifically, the stream interface  3703  demultiplexes the various data included in the multiplexed data demodulated by the demodulator  3702 , such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller such as a CPU. The stream interface  3703  then extracts and multiplexes only the indicated demultiplexed data, thus generating new multiplexed data. The data to be extracted from the demultiplexed data may be determined by the user or may be determined in advance according to the type of stream output interface  3709 . 
     According to this structure, the receiver  3700  is able to extract and output only the required data to an external device. As such, fewer multiplexed data are output using less communication bandwidth. 
     Although the above explanation describes the stream output interface  3709  as outputting multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, the video data included in the multiplexed data so obtained may be converted by using a different video coding method than the original video coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The stream output interface  3709  may then output the converted video data as new multiplexed data. Here, the video coding method used to generate the new video data may conform to a different standard than that used to generate the original video data. Alternatively, the same video coding method may be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding may be converted by using a different audio coding method than the original audio coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The stream output interface  3709  may then output the converted audio data as new multiplexed data. 
     Here, the process by which the audio or video data included in the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding is converted so as to reduce the amount of data or the bit rate thereof is performed by, for example, the stream interface  3703  or the signal processor  3704 . Specifically, the stream interface  3703  demultiplexes the various data included in the multiplexed data demodulated by the demodulator  3702 , such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller. The signal processor  3704  then performs processing to convert the video data so demultiplexed by using a different video coding method than the original video coding method applied thereto, and performs processing to convert the audio data so demultiplexed by using a different video coding method than the original audio coding method applied thereto. As instructed by the controller, the stream interface  3703  then multiplexes the converted audio and video data, thus generating new multiplexed data. The signal processor  3704  may, in accordance with instructions from the controller, performing conversion processing on either the video data or the audio data, alone, or may perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion may be specified by the user or determined in advance according to the type of stream output interface  3709 . 
     According to this structure, the receiver  3700  is able to modify the bit rate of the video and audio data for output according to the speed of communication with the external device. Thus, despite the speed of communication with an external device being slower than the bit rate of the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding, by outputting new multiplexed data from the stream output interface to the external device, the user is able to use the new multiplexed data with other communication devices. 
     The receiver  3700  further includes an audiovisual output interface  3711  that outputs audio and video signals decoded by the signal processor  3704  to the external device through an external communications medium. The audiovisual output interface  3711  may be, for example, a wireless communication device transmitting modulated audiovisual data to an external device using a wireless transmission method conforming to a wireless communication standard such as Wi-Fi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD, Bluetooth™, ZigBee™, and so on through a wireless medium. The stream output interface  3709  may also be a wired communication device transmitting modulated audiovisual data to an external device using a communication method conforming to a wired communication standard such as Ethernet™, USB, PLC, HDMI, and so on through a wired transmission path connected to the stream output interface  3709 . Furthermore, the stream output interface  3709  may be a terminal for connecting a cable that outputs analogue audio signals and video signals as-is. 
     According to such a structure, the user is able to use the audio signals and video signals decoded by the signal processor  3704  with an external device. 
     Further, the receiver  3700  includes an operation input unit  3710  that receives user operations as input. The receiver  3700  behaves in accordance with control signals input by the operation input unit  3710  according to user operations, such as by switching the power supply ON or OFF, changing the channel being received, switching subtitle display ON or OFF, switching between languages, changing the volume output by the audio output unit  3706 , and various other operations, including modifying the settings for receivable channels and the like. 
     The receiver  3700  may further include functionality for displaying an antenna level representing the received signal quality while the receiver  3700  is receiving a signal. The antenna level may be, for example, a index displaying the received signal quality calculated according to the RSSI (Received Signal Strength Indicator), the received signal magnetic field strength, the C/N (carrier-to-noise) ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on, received by the receiver  3700  and indicating the level and the quality of a received signal. In such circumstances, the demodulator  3702  includes a signal quality calibrator that measures the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on. In response to user operations, the receiver  3700  displays the antenna level (signal level, signal quality) in a user-recognizable format on the video display unit  3707 . The display format for the antenna level (signal level, signal quality) may be a numerical value displayed according to the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on, or may be an image display that varies according to the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on. The receiver  3700  may display multiple antenna level (signal level, signal quality) calculated for each stream s 1 , s 2 , and so on demultiplexed using the reception method discussed in the above-described Embodiments, or may display a single antenna level (signal level, signal quality) calculated for all such streams. When the video data and audio data composing a program are transmitted hierarchically, the signal level (signal quality) may also be displayed for each hierarchical level. 
     According to the above structure, the user is given an understanding of the antenna level (signal level, signal quality) numerically or visually during reception using the reception methods discussed in the above-described Embodiments. 
     Although the above example describes the receiver  3700  as including the audio output unit  3706 , the video display unit  3707 , the drive  3708 , the stream output interface  3709 , and the audiovisual output interface  3711 , all of these components are not strictly necessary. As long as the receiver  3700  includes at least one of the above-described components, the user is able to use the multiplexed data obtained through demodulation by the demodulator  3702  and error-correcting decoding. Any receiver may be freely combined with the above-described components according to the usage method. 
     (Multiplexed Data) 
     The following is a detailed description of a sample configuration of multiplexed data. The data configuration typically used in broadcasting is an MPEG-2 transport stream (TS). Therefore the following description describes an example related to MPEG2-TS. However, the data configuration of the multiplexed data transmitted by the transmission and reception methods discussed in the above-described Embodiments is not limited to MPEG2-TS. The advantageous effects of the above-described Embodiments are also achievable using any other data structure. 
       FIG.  38    illustrates a sample configuration for multiplexed data. As shown, the multiplexed data are elements making up programmes (or events, being a portion thereof) currently provided by various services. For example, one or more video streams, audio streams, presentation graphics (PG) streams, interactive graphics (IG) streams, and other such element streams are multiplexed to obtain the multiplexed data. When a broadcast program provided by the multiplexed data is a movie, the video streams represent main video and sub video of the movie, the audio streams represent main audio of the movie and sub-audio to be mixed with the main audio, and the presentation graphics streams represent subtitles for the movie. Main video refers to video images normally presented on a screen, whereas sub-video refers to video images (for example, images of text explaining the outline of the movie) to be presented in a small window inserted within the video images. The interactive graphics streams represent an interactive display made up of GUI (Graphical User Interface) components presented on a screen. 
     Each stream included in the multiplexed data is identified by an identifier, termed a PID, uniquely assigned to the stream. For example, PID 0x1011 is assigned to the video stream used for the main video of the movie, PIDs 0x1100 through 0x111F are assigned to the audio streams, PIDs 0x1200 through 0x121F are assigned to the presentation graphics, PIDs 0x1400 through 0x141F are assigned to the interactive graphics, PIDs 0x1B00 through 0x1B1F are assigned to the video streams used for the sub-video of the movie, and PIDs 0x1A00 through 0x1A1F are assigned to the audio streams used as sub-audio to be mixed with the main audio of the movie. 
       FIG.  39    is a schematic diagram illustrating an example of the multiplexed data being multiplexed. First, a video stream  3901 , made up of a plurality of frames, and an audio stream  3904 , made up of a plurality of audio frames, are respectively converted into PES packet sequence  3902  and  3905 , then further converted into TS packets  3903  and  3906 . Similarly, a presentation graphics stream  3911  and an interactive graphics stream  3914  are respectively converted into PES packet sequence  3912  and  3915 , then further converted into TS packets  3913  and  3916 . The multiplexed data  3917  is made up of the TS packets  3903 ,  3906 ,  3913 , and  3916  multiplexed into a single stream. 
       FIG.  40    illustrates further details of a PES packet sequence as contained in the video stream. The first tier of  FIG.  40    shows a video frame sequence in the video stream. The second tier shows a PES packet sequence. Arrows yy1, yy2, yy3, and yy4 indicate the plurality of Video Presentation Units, which are I-pictures, B-pictures, and P-pictures, in the video stream as divided and individually stored as the payload of a PES packet. Each PES packet has a PES header. A PES header contains a PTS (Presentation Time Stamp) at which the picture is to be displayed, a DTS (Decoding Time Stamp) at which the picture is to be decoded, and so on. 
       FIG.  41    illustrates the structure of a TS packet as ultimately written into the multiplexed data. A TS packet is a 188-byte fixed-length packet made up of a 4-byte PID identifying the stream and of a 184-byte TS payload containing the data. The above-described PES packets are divided and individually stored as the TS payload. For a BD-ROM, each TS packet has a 4-byte TP_Extra_Header affixed thereto to build a 192-byte source packet, which is to be written as the multiplexed data. The TP_Extra_Header contains information such as an Arrival_Time_Stamp (ATS). The ATS indicates a time for starring transfer of the TS packet to the PID filter of a decoder. The multiplexed data are made up of source packets arranged as indicated in the bottom tier of  FIG.  41   . A SPN (source packet number) is incremented for each packet, beginning at the head of the multiplexed data. 
     In addition to the video streams, audio streams, presentation graphics streams, and the like, the TS packets included in the multiplexed data also include a PAT (Program Association Table), a PMT (Program Map Table), a PCR (Program Clock Reference) and so on. The PAT indicates the PID of a PMT used in the multiplexed data, and the PID of the PAT itself is registered as 0. The PMT includes PIDs identifying the respective streams, such as video, audio and subtitles, contained in the multiplexed data and attribute information (frame rate, aspect ratio, and the like) of the streams identified by the respective PIDs. In addition, the PMT includes various types of descriptors relating to the multiplexed data. One such descriptor may be copy control information indicating whether or not copying of the multiplexed data is permitted. The PCR includes information for synchronizing the ATC (Arrival Time Clock) serving as the chronological axis of the ATS to the STC (System Time Clock) serving as the chronological axis of the PTS and DTS. Each PCR packet includes an STC time corresponding to the ATS at which the packet is to be transferred to the decoder. 
       FIG.  42    illustrates the detailed data configuration of a PMT. The PMT starts with a PMT header indicating the length of the data contained in the PMT. Following the PMT header, descriptors pertaining to the multiplexed data are arranged. One example of a descriptor included in the PMT is the copy control information described above. Following the descriptors, stream information pertaining to the respective streams included in the multiplexed data is arranged. Each piece of stream information is composed of stream descriptors indicating a stream type identifying a compression codec employed for a corresponding stream, a PID for the stream, and attribute information (frame rate, aspect ratio, and the like) of the stream. The PMT includes the same number of stream descriptors as the number of streams included in the multiplexed data. 
     When recorded onto a recoding medium or the like, the multiplexed data are recorded along with a multiplexed data information file. 
       FIG.  43    illustrates a sample configuration for the multiplexed data information file. As shown, the multiplexed data information file is management information for the multiplexed data, is provided in one-to-one correspondence with the multiplexed data, and is made up of multiplexed data information, stream attribute information, and an entry map. 
     The multiplexed data information is made up of a system rate, a playback start time, and a playback end time. The system rate indicates the maximum transfer rate of the multiplexed data to the PID filter of a later-described system target decoder. The multiplexed data includes ATS at an interval set so as not to exceed the system rate. The playback start time is set to the time specified by the PTS of the first video frame in the multiplexed data, whereas the playback end time is set to the time calculated by adding the playback duration of one frame to the PTS of the last video frame in the multiplexed data. 
       FIG.  44    illustrates a sample configuration for the stream attribute information included in the multiplexed data information file. As shown, the stream attribute information is attribute information for each stream included in the multiplexed data, registered for each PID. That is, different pieces of attribute information are provided for different streams, namely for the video streams, the audio streams, the presentation graphics streams, and the interactive graphics streams. The video stream attribute information indicates the compression codec employed to compress the video stream, the resolution of individual pictures constituting the video stream, the aspect ratio, the frame rate, and so on. The audio stream attribute information indicates the compression codec employed to compress the audio stream, the number of channels included in the audio stream, the language of the audio stream, the sampling frequency, and so on. This information is used to initialize the decoder before playback by a player. 
     In the present Embodiment, the stream type included in the PMT is used among the information included in the multiplexed data. When the multiplexed data are recorded on a recording medium, the video stream attribute information included in the multiplexed data information file is used. Specifically, the video coding method and device described in any of the above Embodiments may be modified to additionally include a step or unit of setting a specific piece of information in the stream type included in the PMT or in the video stream attribute information. The specific piece of information is for indicating that the video data are generated by the video coding method and device described in the Embodiment. According to such a structure, video data generated by the video coding method and device described in any of the above Embodiments is distinguishable from video data compliant with other standards. 
       FIG.  45    illustrates a sample configuration of an audiovisual output device  4500  that includes a reception device  4504  receiving a modulated signal that includes audio and video data transmitted by a broadcaster (base station) or data intended for broadcasting. The configuration of the reception device  4504  corresponds to the reception device  3700  from  FIG.  37   . The audiovisual output device  4500  incorporates, for example, an OS (Operating System), or incorporates a communication device  4506  for connecting to the Internet (e.g., a communication device intended for a wireless LAN (Local Area Network) or for Ethernet™). As such, a video display unit  4501  is able to simultaneously display audio and video data, or video in video data for broadcast  4502 , and hypertext  4503  (from the World Wide Web) provided over the Internet. By operating a remote control  4507  (alternatively, a mobile phone or keyboard), either of the video in video data for broadcast  4502  and the hypertext  4503  provided over the Internet may be selected to change operations. For example, when the hypertext  4503  provided over the Internet is selected, the website displayed may be changed by remote control operations. When audio and video data, or video in video data for broadcast  4502  is selected, information from a selected channel (selected (television) program or audio broadcast) may be transmitted by the remote control  4507 . As such, an interface  4505  obtains the information transmitted by the remote control. The reception device  4504  performs processing such as demodulation and error-correction corresponding to the selected channel, thereby obtaining the received data. At this point, the reception device  4504  obtains control symbol information that includes information on the transmission method (as described using  FIG.  5   ) from control symbols included the signal corresponding to the selected channel. As such, the reception device  4504  is able to correctly set the reception operations, demodulation scheme, error-correction method and so on, thus enabling the data included in the data symbols transmitted by the broadcaster (base station) to be obtained. Although the above description is given for an example of the user using the remote control  4507 , the same operations apply when the user presses a selection key embedded in the audiovisual output device  4500  to select a channel. 
     In addition, the audiovisual output device  4500  may be operated using the Internet. For example, the audiovisual output device  4500  may be made to record (store) a program through another terminal connected to the Internet. (Accordingly, the audiovisual output device  4500  should include the drive  3708  from  FIG.  37   .) The channel is selected before recording begins. As such, the reception device  4504  performs processing such as demodulation and error-correction corresponding to the selected channel, thereby obtaining the received data. At this point, the reception device  4504  obtains control symbol information that includes information on the transmission method (the transmission method, modulation scheme, error-correction method, and so on from the above-described Embodiments) (as described using  FIG.  5   ) from control symbols included the signal corresponding to the selected channel. As such, the reception device  4504  is able to correctly set the reception operations, demodulation scheme, error-correction method and so on, thus enabling the data included in the data symbols transmitted by the broadcaster (base station) to be obtained. 
     (Supplement) 
     The present description considers a communications/broadcasting device such as a broadcaster, a base station, an access point, a terminal, a mobile phone, or the like provided with the transmission device, and a communications device such as a television, radio, terminal, personal computer, mobile phone, access point, base station, or the like provided with the reception device. The transmission device and the reception device pertaining to the present invention are communication devices in a form able to execute applications, such as a television, radio, personal computer, mobile phone, or similar, through connection to some sort of interface (e.g., USB). 
     Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (namely preamble, unique word, postamble, reference symbols, scattered pilot symbols and so on), symbols intended for control information, and so on may be freely arranged within the frame. Although pilot symbols and symbols intended for control information are presently named, such symbols may be freely named otherwise as the function thereof remains the important consideration. 
     Provided that a pilot symbol, for example, is a known symbol modulated with PSK modulation in the transmitter and receiver (alternatively, the receiver may be synchronized such that the receiver knows the symbols transmitted by the transmitter), the receiver is able to use this symbol for frequency synchronization, time synchronization, channel estimation (CSI (Channel State Information) estimation for each modulated signal), signal detection, and the like. 
     The symbols intended for control information are symbols transmitting information (such as the modulation scheme, error-correcting coding method, coding rate of error-correcting codes, and setting information for the top layer used in communications) that is transmitted to the receiving party in order to execute transmission of non-data (i.e., applications). 
     The present invention is not limited to the Embodiments, but may also be realized in various other ways. For example, while the above Embodiments describe communication devices, the present invention is not limited to such devices and may be implemented as software for the corresponding communications method. 
     Although the above-described Embodiments describe phase changing methods for methods of transmitting two modulated signals from two antennas, no limitation is intended in this regard. Precoding and a change of phase may be performed on four signals that have been mapped to generate four modulated signals transmitted using four antennas. That is, the present invention is applicable to performing a change of phase on N signals that have been mapped and precoded to generate N modulated signals transmitted using N antennas. 
     Although the above-described Embodiments describe examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communications system, the present invention is not limited in this regard and is also applicable to MISO (Multiple Input Single Output) communications systems. In a MISO system, the reception device does not include antenna  701 _Y, wireless unit  703 _Y, channel fluctuation estimator  707 _ 1  for modulated signal z 1 , and channel fluctuation estimator  707 _ 2  for modulated signal z 2  from  FIG.  7   . However, the processing described in Embodiment 1 may still be executed to estimate r 1  and r 2 . Technology for receiving and decoding a plurality of signals transmitted simultaneously at a common frequency are received by a single antenna is widely known. The present invention is additional processing supplementing conventional technology for a signal processor reverting a phase changed by the transmitter. 
     Although the present invention describes examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communications system, the present invention is not limited in this regard and is also applicable to MISO systems. In a MISO system, the transmission device performs precoding and change of phase such that the points described thus far are applicable. However, the reception device does not include antenna  701 _Y, wireless unit  703 _Y, channel fluctuation estimator  707 _ 1  for modulated signal z 1 , and channel fluctuation estimator  707 _ 2  for modulated signal z 2  from  FIG.  7   . However, the processing described in the present description may still be executed to estimate the data transmitted by the transmission device. Technology for receiving and decoding a plurality of signals transmitted simultaneously at a common frequency are received by a single antenna is widely known (a single-antenna receiver may apply ML operations (Max-log APP or similar)). The present invention may have the signal processor  711  from  FIG.  7    perform demodulation (detection) by taking the precoding and change of phase applied by the transmitter into consideration. 
     The present description uses terms such as precoding, precoding weights, precoding matrix, and so on. The terminology itself may be otherwise (e.g., may be alternatively termed a codebook) as the key point of the present invention is the signal processing itself. 
     Furthermore, although the present description discusses examples mainly using OFDM as the transmission method, the invention is not limited in this manner. Multi-carrier methods other than OFDM and single-carrier methods may all be used to achieve similar Embodiments. Here, spread-spectrum communications may also be used. When single-carrier methods are used, the change of phase is performed with respect to the time domain. 
     In addition, although the present description discusses the use of ML operations, APP, Max-log APP, ZF, MMSE and so on by the reception device, these operations may all be generalized as wave detection, demodulation, detection, estimation, and demultiplexing as the soft decision results (log-likelihood and log-likelihood ratio) and the hard decision results (zeroes and ones) obtained thereby are the individual bits of data transmitted by the transmission device. 
     Different data may be transmitted by each stream s 1 ( t ) and s 2 ( t ) (s 1 ( i ), s 2 ( i )), or identical data may be transmitted thereby. 
     The two stream baseband signals s 1 ( i ) and s 2 ( i ) (where i indicates sequence (with respect to time or (carrier) frequency)) undergo precoding and a regular change of phase (the order of operations may be freely reversed) to generate two post-processing baseband signals z 1 ( i ) and z 2 ( i ). For post-processing baseband signal z 1 ( i ), the in-phase component I is I 1 (i) while the quadrature component is Q 1 (i), and for post processing baseband signal z 2 ( i ), the in-phase component is I 1 (i) while the quadrature component is Q 2 (i). The baseband components may be switched, as long as the following holds. 
     Let the in-phase component and the quadrature component of switched baseband signal r 1 ( i ) be I 1 (i) and Q 2 (i), and the in-phase component and the quadrature component of switched baseband signal r 2 ( i ) be I 2 (i) and Q 1 (i). 
     The modulated signal corresponding to switched baseband signal r 1 ( i ) is transmitted by transmit antenna  1  and the modulated signal corresponding to switched baseband signal r 2 ( i ) is transmitted from transmit antenna  2 , simultaneously on a common frequency. As such, the modulated signal corresponding to switched baseband signal r 1 ( i ) and the modulated signal corresponding to switched baseband signal r 2 ( i ) are transmitted from different antennas, simultaneously on a common frequency. Alternatively,
         For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i) while the quadrature component may be I 2 (i), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i) while the quadrature component may be Q 2 (i).   For switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i) while the quadrature component may be I 1 (i), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i) while the quadrature component may be Q 2 (i).   For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i) while the quadrature component may be I 2 (i), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i) while the quadrature component may be Q 1 (i).   For switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i) while the quadrature component may be I 1 (i), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i) while the quadrature component may be Q 1 (i).   For switched baseband signal r 1 ( i ), the in-phase component may be I 1 ( i ) while the quadrature component may be Q 2 ( i ), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 ( i ) while the quadrature component may be I 2 ( i ).   For switched baseband signal r 1 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 2 ( i ), the in-phase component may be I 2 ( i ) while the quadrature component may be Q 1 ( i ).   For switched baseband signal r 1 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 ( i ) while the quadrature component may be I 2 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 ( i ) while the quadrature component may be I 2 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 ( i ) while the quadrature component may be Q 2 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be I 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 ( i ) while the quadrature component may be Q 2 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 ( i ) while the quadrature component may be I 2 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be Q 1 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be I 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be Q 1 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 ( i ) while the quadrature component may be Q 2 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be I 2 ( i ) while the quadrature component may be Q 1 (i).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 (i) while the quadrature component may be Q 2 (i), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 (i) while the quadrature component may be I 2 (i).   For switched baseband signal r 2 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be I 2 ( i ) while the quadrature component may be Q 1 ( i ).   For switched baseband signal r 2 ( i ), the in-phase component may be Q 2 ( i ) while the quadrature component may be I 1 ( i ), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 ( i ) while the quadrature component may be I 2 ( i ).       

     Alternatively, although the above description discusses performing two types of signal processing on both stream signals so as to switch the in-phase component and quadrature component of the two signals, the invention is not limited in this manner. The two types of signal processing may be performed on more than two streams, so as to switch the in-phase component and quadrature component thereof. 
     Alter, while the above examples describe switching performed on baseband signals having a common timestamp (common (sub-)carrier) frequency), the baseband signals being switched need not necessarily have a common timestamp (common (sub-)carrier) frequency). For example, any of the following are possible.
         For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be I 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be Q 2 (i+w).   For switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i+v) while the quadrature component may be Q 1 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be Q 2 (i+w).   For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i+v) while the quadrature component may be Q 1 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 1 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be I 2 (i+w).   For switched baseband signal r 1 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be I 1 (i+v), and for switched baseband signal r 2 ( i ), the in-phase component may be I 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 1 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be I 1 (i+v), and for switched baseband signal r 2 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be I 2 (i+w).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be Q 2 (i+w).   For switched baseband signal r 2 ( i ), the in-phase component may be I 2 (i+v) while the quadrature component may be Q 1 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be Q 2 (i+w).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 2 ( i ), the in-phase component may be I 2 (i+v) while the quadrature component may be Q 1 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 2 ( i ), the in-phase component may be I 1 (i+v) while the quadrature component may be Q 2 (i+w), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be I 2 (i+w).   For switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be I 1 (i+v), and for switched baseband signal r 1 ( i ), the in-phase component may be I 2 (i+w) while the quadrature component may be Q 1 (i+v).   For switched baseband signal r 2 ( i ), the in-phase component may be Q 2 (i+w) while the quadrature component may be I 1 (i+v), and for switched baseband signal r 1 ( i ), the in-phase component may be Q 1 (i+v) while the quadrature component may be I 2 (i+w).       

       FIG.  55    illustrates a baseband signal switcher  5502  explaining the above. As shown, of the two processed baseband signals z 1 ( i )  5501 _ 1  and z 2 ( i )  5501 _ 2 , processed baseband signal z 1 ( i )  5501 _ 1  has in-phase component I 1 ( i ) and quadrature component Q 1 ( i ), while processed baseband signal z 2 ( i )  5501 _ 2  has in-phase component I 2 ( i ) and quadrature component Q 2 ( i ). Then, after switching, switched baseband signal r 1 ( i )  5503 _ 1  has in-phase component I r1 ( i ) and quadrature component Q r1 ( i ), while switched baseband signal r 2 ( i )  5503 _ 2  has in-phase component I r2 ( i ) and quadrature component Q r2 ( i ). The in-phase component I r1 ( i ) and quadrature component Q r1 ( i ) of switched baseband signal r 1 ( i )  5503 _ 1  and the in-phase component I r2 ( i ) and quadrature component Q r2 ( i ) of switched baseband signal r 2 ( i )  5503 _ 2  may be expressed as any of the above. Although this example describes switching performed on baseband signals having a common timestamp (common ((sub-)carrier) frequency) and having undergone two types of signal processing, the same may be applied to baseband signals having undergone two types of signal processing but having different timestamps (different ((sub-)carrier) frequencies). 
     Each of the transmit antennas of the transmission device and each of the receive antennas of the reception device shown in the figures may be formed by a plurality of antennas. 
     The present description uses the symbol V, which is the universal quantifier, and the symbol ∃, which is the existential quantifier. 
     Furthermore, the present description uses the radian as the unit of phase in the complex plane, e.g., for the argument thereof. 
     When dealing with the complex plane, the coordinates of complex numbers are expressible by way of polar coordinates. For a complex number z=a+jb (where a and b are real numbers and j is the imaginary unit), the corresponding point (a, b) on the complex plane is expressed with the polar coordinates [r, θ], converted as follows:
 
α= r ×cos θ
 
 b=r ×sin θ
 
[Math. 49]
 
 r =√{square root over ( a   2   +b   2 )}  (formula 49)
 
     where r is the absolute value of z (r=|z|), and θ is the argument thereof. As such, z=a+jb is expressible as re jθ . 
     In the present invention, the baseband signals s 1 , s 2 , z 1 , and z 2  are described as being complex signals. A complex signal made up of in-phase signal I and quadrature signal Q is also expressible as complex signal I+jQ. Here, either of I and Q may be equal to zero. 
       FIG.  46    illustrates a sample broadcasting system using the phase changing method described in the present description. As shown, a video encoder  4601  takes video as input, performs video encoding, and outputs encoded video data  4602 . An audio encoder  4603  takes audio as input, performs audio encoding, and outputs encoded audio data  4604 . A data encoder  4605  takes data as input, performs data encoding (e.g., data compression), and outputs encoded data  4606 . Taken as a whole, these components form a source information encoder  4600 . 
     A transmitter  4607  takes the encoded video data  4602 , the encoded audio data  4604 , and the encoded data  4606  as input, performs error-correcting coding, modulation, precoding, and phase changing (e.g., the signal processing by the transmission device from  FIG.  3   ) on a subset of or on the entirety of these, and outputs transmit signals  4608 _ 1  through  4608 _N. Transmit signals  4608 _ 1  through  4608 _N are then transmitted by antennas  4609 _ 1  through  4609 _N as radio waves. 
     A receiver  4612  takes received signals  4611 _ 1  through  4611 _M received by antennas  4610 _ 1  through  4610 _M as input, performs processing such as frequency conversion, change of phase, decoding of the precoding, log-likelihood ratio calculation, and error-correcting decoding (e.g., the processing by the reception device from  FIG.  7   ), and outputs received data  4613 ,  4615 , and  4617 . A source information decoder  4619  takes the received data  4613 ,  4615 , and  4617  as input. A video decoder  4614  takes received data  4613  as input, performs video decoding, and outputs a video signal. The video is then displayed on a television display. An audio decoder  4616  takes received data  4615  as input. The audio decoder  4616  performs audio decoding and outputs an audio signal. the audio is then played through speakers. A data decoder  4618  takes received data  4617  as input, performs data decoding, and outputs information. 
     In the above-described Embodiments pertaining to the present invention, the number of encoders in the transmission device using a multi-carrier transmission method such as OFDM may be any number, as described above. Therefore, as in  FIG.  4   , for example, the transmission device may have only one encoder and apply a method of distributing output to the multi-carrier transmission method such as OFDM. In such circumstances, the wireless units  310 A and  310 B from  FIG.  4    should replace the OFDM-related processors  1301 A and  1301 B from  FIG.  12   . The description of the OFDM-related processors is as given for Embodiment 1. 
     Although Embodiment 1 gives Math. 36 (formula 36) as an example of a precoding matrix, another precoding matrix may also be used, when the following method is applied. 
     
       
         
           
             
               
                 
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     In the precoding matrices of Math. 36 (formula 36) and Math. 50 (formula 50), the value of α is set as given by Math. 37 (formula 37) and Math. 38 (formula 38). However, no limitation is intended in this manner. A simple precoding matrix is obtainable by setting α=1, which is also a valid value. 
     In Embodiment A1, the phase changers from  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51   , and  53  are indicated as having a phase changing value of PHASE[i] (where i=0, 1, 2, . . . , N−2, N−1 (i being an integer between 0 and N−1)) to achieve a period (cycle) of N (value reached given that  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    perform a change of phase on only one baseband signal). The present description discusses performing a change of phase on one precoded baseband signal (i.e., in  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51  and  53   ) namely on precoded baseband signal z 2 ′. Here, PHASE[k] is calculated as follows. 
     
       
         
           
             
               
                 
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     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). When N=5, 7, 9, 11, or 15, the reception device is able to obtain good data reception quality. 
     Although the present description discusses the details of phase changing methods involving two modulated signals transmitted by a plurality of antennas, no limitation is intended in this regard. Precoding and a change of phase may be performed on three or more baseband signals on which mapping has been performed according to a modulation scheme, followed by predetermined processing on the post-phase change baseband signals and transmission using a plurality of antennas, to realize the same results. 
     Programs for executing the above transmission method may, for example, be stored in advance in ROM (Read-Only Memory) and be read out for operation by a CPU. 
     Furthermore, the programs for executing the above transmission method may be stored on a computer-readable recording medium, the programs stored in the recording medium may be loaded in the RAM (Random Access Memory) of the computer, and the computer may be operated in accordance with the programs. 
     The components of the above-described Embodiments may be typically assembled as an LSI (Large Scale Integration), a type of integrated circuit. Individual components may respectively be made into discrete chips, or a subset or entirety of the components may be made into a single chip. Although an LSI is mentioned above, the terms IC (Integrated Circuit), system LSI, super LSI, or ultra LSI may also apply, depending on the degree of integration. Furthermore, the method of integrated circuit assembly is not limited to LSI. A dedicated circuit or a general-purpose processor may be used. After LSI assembly, a FPGA (Field Programmable Gate Array) or reconfigurable processor may be used. 
     Furthermore, should progress in the field of semiconductors or emerging technologies lead to replacement of LSI with other integrated circuit methods, then such technology may of course be used to integrate the functional blocks. Applications to biotechnology are also plausible. 
     Embodiment C1 
     Embodiment 1 explained that the precoding matrix in use may be switched when transmission parameters change. The present Embodiment describes a detailed example of such a case, where, as described above (in the supplement), the transmission parameters change such that streams s 1 ( t ) and s 2 ( t ) switch between transmitting different data and transmitting identical data, and the precoding matrix and phase changing method being used are switched accordingly. 
     The example of the present Embodiment describes a situation where two modulated signals transmitted from two different transmit antenna alternate between having the modulated signals include identical data and having the modulated signals each include different data. 
       FIG.  56    illustrates a sample configuration of a transmission device switching between transmission methods, as described above. In  FIG.  56   , components operating in the manner described for  FIG.  54    use identical reference numbers. As shown,  FIG.  56    differs from  FIG.  54    in that a distributor  404  takes the frame configuration signal  313  as input. The operations of the distributor  404  are described using  FIG.  57   . 
       FIG.  57    illustrates the operations of the distributor  404  when transmitting identical data and when transmitting different data. As shown, given encoded data x 1 , x 2 , x 3 , x 4 , x 5 , x 6 , and so on, when transmitting identical data, distributed data  405  is given as x 1 , x 2 , x 3 , x 4 , x 5 , x 6 , and so on, while distributed data  405 B is similarly given as x 1 , x 2 , x 3 , x 4 , x 5 , x 6 , and so on. 
     On the other hand, when transmitting different data, distributed data  405 A are given as x 1 , x 3 , x 5 , x 7 , x 9 , and so on, while distributed data  405 B are given as x 2 , x 4 , x 6 , x 8 , x 10 , and so on. 
     The distributor  404  determines, according to the frame configuration signal  313  taken as input, whether the transmission mode is identical data transmission or different data transmission. 
     An alternative method to the above is shown in  FIG.  58   . As shown, when transmitting identical data, the distributor  404  outputs distributed data  405 A as x 1 , x 2 , x 3 , x 4 , x 5 , x 6 , and so on, while outputting nothing as distributed data  405 B. Accordingly, when the frame configuration signal  313  indicates identical data transmission, the distributor  404  operates as described above, while interleaver  304 B and mapper  306 B from  FIG.  56    do not operate. Thus, only baseband signal  307 A output by mapper  306 A from  FIG.  56    is valid, and is taken as input by both weighting unit  308 A and  308 B. 
     One characteristic feature of the present Embodiment is that, when the transmission mode switches from identical data transmission to different data transmission, the precoding matrix may also be switched. As indicated by Math. 36 (formula 36) and Math. 39 (formula 39) in Embodiment 1, given a matrix made up of w 11 , w 12 , w 21 , and w 22 , the precoding matrix used to transmit identical data may be as follows. 
     
       
         
           
             
               
                 
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                             ⁢ 
                             2 
                           
                         
                       
                       
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             2 
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     ( 
                     
                       
                         
                           a 
                         
                         
                           0 
                         
                       
                       
                         
                           0 
                         
                         
                           a 
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     52 
                   
                   ) 
                 
               
             
           
         
       
     
     where a is a real number (a may also be a complex number, but given that the baseband signal input as a result of precoding undergoes a change of phase, a real number is preferable for considerations of circuit size and complexity reduction). Also, when a is equal to one, the weighting units  308 A and  308 B do not perform weighting and output the input signal as-is. 
     Accordingly, when transmitting identical data, the weighted baseband signals  309 A and  316 B are identical signals output by the weighting units  308 A and  308 B. 
     When the frame configuration signal  313  indicates identical transmission mode, a phase changer  5201  performs a change of phase on weighted baseband signal  309 A and outputs post-phase change baseband signal  5202 . Similarly, when the frame configuration signal indicates identical transmission mode, phase changer  317 B performs a change of phase on weighted baseband signal  316 B and outputs post-phase change baseband signal  309 B. The change of phase performed by phase changer  5201  is of e jA(t)  (alternatively, e jA(f)  or e jA(t,f) ) (where t is time and f is frequency) (accordingly, e jA(t)  (alternatively, e jA(f)  or e jA(t,f) ) is the value by which the input baseband signal is multiplied), and the change of phase performed by phase changer  317 B is of e jB(t)  (alternatively, e jB(f)  or e jB(t,f) ) (where t is time and f is frequency) (accordingly, e jB(t)  (alternatively, e jB(f)  or e jB(t,f) ) is the value by which the input baseband signal is multiplied). As such, the following condition is satisfied. 
     [Math. 53] 
     Some time t satisfies
 
 e   jA(   t ) ≠e   jB(t)  
 
     (Or, some (carrier) frequency f satisfies e jA(f) ≠e jB(f) ) 
     (Or, some (carrier) frequency f and time t satisfy e jA(t,f) ≠e jB (t,f) ) 
     As such, the transmit signal is able to reduce multi-path influence and thereby improve data reception quality for the reception device. (However, the change of phase may also be performed by only one of the weighted baseband signals  309 A and  316 B.) 
     In  FIG.  56   , when OFDM is used, processing such as IFFT and frequency conversion is performed on post-phase change baseband signal  5202 , and the result is transmitted by a transmit antenna. (See  FIG.  13   ) (Accordingly, post-phase change baseband signal  5202  may be considered the same as signal  1301 A from  FIG.  13   .) Similarly, when OFDM is used, processing such as IFFT and frequency conversion is performed on post-phase change baseband signal  309 B, and the result is transmitted by a transmit antenna. (See  FIG.  13   ) (Accordingly, post-phase change baseband signal  309 B may be considered the same as signal  1301 B from  FIG.  13   .) 
     When the selected transmission mode indicates different data transmission, then any of Math. 36 (formula 36), Math. 39 (formula 39), and Math. 50 (formula 50) given in Embodiment 1 may apply. Significantly, the phase changers  5201  and  317 B from  FIG.  56    use a different phase changing method than when transmitting identical data. Specifically, as described in Embodiment 1, for example, phase changer  5201  performs the change of phase while phase changer  317 B does not, or phase changer  317 B performs the change of phase while phase changer  5201  does not. Only one of the two phase changers performs the change of phase. As such, the reception device obtains good data reception quality in the LOS environment as well as the NLOS environment. 
     When the selected transmission mode indicates different data transmission, the precoding matrix may be as given in Math. 52 (formula 52), or as given in any of Math. 36 (formula 36), Math. 50 (formula 50), and Math. 39 (formula 39), or may be a precoding matrix unlike that given in Math. 52 (formula 52). Thus, the reception device is especially likely to experience improvements to data reception quality in the LOS environment. 
     Furthermore, although the present Embodiment discusses examples using OFDM as the transmission method, the invention is not limited in this manner. Multi-carrier methods other than OFDM and single-carrier methods may all be used to achieve similar Embodiments. Here, spread-spectrum communications may also be used. When single-carrier methods are used, the change of phase is performed with respect to the time domain. 
     As explained in Embodiment 3, when the transmission method involves different data transmission, the change of phase is carried out on the data symbols, only. However, as described in the present Embodiment, when the transmission method involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.) 
     Embodiment C2 
     The present Embodiment describes a configuration method for a base station corresponding to Embodiment C1. 
       FIG.  59    illustrates the relationship of a base stations (broadcasters) to terminals. A terminal P ( 5907 ) receives transmit signal  5903 A transmitted by antenna  5904 A and transmit signal  5905 A transmitted by antenna  5906 A of broadcaster A ( 5902 A), then performs predetermined processing thereon to obtained received data. 
     A terminal Q ( 5908 ) receives transmit signal  5903 A transmitted by antenna  5904 A of base station A ( 5902 A) and transmit signal  593 B transmitted by antenna  5904 B of base station B ( 5902 B), then performs predetermined processing thereon to obtained received data. 
       FIGS.  60  and  61    illustrate the frequency allocation of base station A ( 5902 A) for transmit signals  5903 A and  5905 A transmitted by antennas  5904 A and  5906 A, and the frequency allocation of base station B ( 5902 B) for transmit signals  5903 B and  5905 B transmitted by antennas  5904 B and  5906 B. In  FIGS.  60  and  61   , frequency is on the horizontal axis and transmission power is on the vertical axis. 
     As shown, transmit signals  5903 A and  5905 A transmitted by base station A ( 5902 A) and transmit signals  5903 B and  5905 B transmitted by base station B ( 5902 B) use at least frequency band X and frequency band Y. Frequency band X is used to transmit data of a first channel, and frequency band Y is used to transmit data of a second channel. 
     Accordingly, terminal P ( 5907 ) receives transmit signal  5903 A transmitted by antenna  5904 A and transmit signal  5905 A transmitted by antenna  5906 A of base station A ( 5902 A), extracts frequency band X therefrom, performs predetermined processing, and thus obtains the data of the first channel. Terminal Q ( 5908 ) receives transmit signal  5903 A transmitted by antenna  5904 A of base station A ( 5902 A) and transmit signal  5903 B transmitted by antenna  5904 B of base station B ( 5902 B), extracts frequency band Y therefrom, performs predetermined processing, and thus obtains the data of the second channel. 
     The following describes the configuration and operations of base station A ( 5902 A) and base station B ( 5902 B). 
     As described in Embodiment C1, both base station A ( 5902 A) and base station B ( 5902 B) incorporate a transmission device configured as illustrated by  FIGS.  56  and  13   . When transmitting as illustrated by  FIG.  60   , base station A ( 5902 A) generates two different modulated signals (on which precoding and a change of phase are performed) with respect to frequency band X as described in Embodiment C1. The two modulated signals are respectively transmitted by the antennas  5904 A and  5906 A. With respect to frequency band Y, base station A ( 5902 A) operates interleaver  304 A, mapper  306 A, weighting unit  308 A, and phase changer from  FIG.  56    to generate modulated signal  5202 . Then, a transmit signal corresponding to modulated signal  5202  is transmitted by antenna  1310 A from  FIG.  13   , i.e., by antenna  5904 A from  FIG.  59   . Similarly, base station B ( 5902 B) operates interleaver  304 A, mapper  306 A, weighting unit  308 A, and phase changer  5201  from  FIG.  56    to generate modulated signal  5202 . Then, a transmit signal corresponding to modulated signal  5202  is transmitted by antenna  1310 A from  FIG.  13   , i.e., by antenna  5904 B from  FIG.  59   . 
     The creation of encoded data in frequency band Y may involve, as shown in  FIG.  56   , generating encoded data in individual base stations, or may involve having one of the base stations generate such encoded data for transmission to other base stations. As an alternative method, one of the base stations may generate modulated signals and be configured to pass the modulated signals so generated to other base stations. 
     Also, in  FIG.  59   , signal  5901  includes information pertaining to the transmission mode (identical data transmission or different data transmission). The base stations obtain this signal and thereby switch between generation methods for the modulated signals in each frequency band. Here, signal  5901  is indicated in  FIG.  59    as being input from another device or from a network. However, configurations where, for example, base station A ( 5902 ) is a master station passing a signal corresponding to signal  5901  to base station B ( 5902 B) are also possible. 
     As explained above, when the base station transmits different data, the precoding matrix and phase changing method are set according to the transmission method to generate modulated signals. 
     On the other hand, to transmit identical data, two base stations respectively generate and transmit modulated signals. In such circumstances, base stations each generating modulated signals for transmission from a common antenna may be considered to be two combined base stations using the precoding matrix given by Math. 52 (formula 52). The phase changing method is as explained in Embodiment C1, for example, and satisfies the conditions of Math. 53 (formula 53). 
     In addition, the transmission method of frequency band X and frequency band Y may vary over time. Accordingly, as illustrated in  FIG.  61   , as time passes, the frequency allocation changes from that indicated in  FIG.  60    to that indicated in  FIG.  61   . 
     According to the present Embodiment, not only can the reception device obtain improved data reception quality for identical data transmission as well as different data transmission, but the transmission devices can also share a phase changer. 
     Furthermore, although the present Embodiment discusses examples using OFDM as the transmission method, the invention is not limited in this manner. Multi-carrier methods other than OFDM and single-carrier methods may all be used to achieve similar Embodiments. Here, spread-spectrum communications may also be used. When single-carrier methods are used, the change of phase is performed with respect to the time domain. 
     As explained in Embodiment 3, when the transmission method involves different data transmission, the change of phase is carried out on the data symbols, only. However, as described in the present Embodiment, when the transmission method involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.) 
     Embodiment C3 
     The present Embodiment describes a configuration method for a repeater corresponding to Embodiment C1. The repeater may also be termed a repeating station. 
       FIG.  62    illustrates the relationship of a base stations (broadcasters) to repeaters and terminals. As shown in  FIG.  63   , base station  6201  at least transmits modulated signals on frequency band X and frequency band Y. Base station  6201  transmits respective modulated signals on antenna  6202 A and antenna  6202 B. The transmission method here used is described later, with reference to  FIG.  63   . 
     Repeater A ( 6203 A) performs processing such as demodulation on received signal  6205 A received by receive antenna  6204 A and on received signal  6207 A received by receive antenna  6206 A, thus obtaining received data. Then, in order to transmit the received data to a terminal, repeater A ( 6203 A) performs transmission processing to generate modulated signals  6209 A and  6211 A for transmission on respective antennas  6210 A and  6212 A. 
     Similarly, repeater B ( 6203 B) performs processing such as demodulation on received signal  6205 B received by receive antenna  6204 B and on received signal  6207 B received by receive antenna  6206 B, thus obtaining received data. Then, in order to transmit the received data to a terminal, repeater B ( 6203 B) performs transmission processing to generate modulated signals  6209 B and  6211 B for transmission on respective antennas  6210 B and  6212 B. Here, repeater B ( 6203 B) is a master repeater that outputs a control signal  6208 . repeater A ( 6203 A) takes the control signal as input. A master repeater is not strictly necessary. Base station  6201  may also transmit individual control signals to repeater A ( 6203 A) and to repeater B ( 6203 B). 
     Terminal P ( 5907 ) receives modulated signals transmitted by repeater A ( 6203 A), thereby obtaining data. Terminal Q ( 5908 ) receives signals transmitted by repeater A ( 6203 A) and by repeater B ( 6203 B), thereby obtaining data. Terminal R ( 6213 ) receives modulated signals transmitted by repeater B ( 6203 B), thereby obtaining data. 
       FIG.  63    illustrates the frequency allocation for a modulated signal transmitted by antenna  6202 A among transmit signals transmitted by the base station, and the frequency allocation of modulated signals transmitted by antenna  6202 B. In  FIG.  63   , frequency is on the horizontal axis and transmission power is on the vertical axis. 
     As shown, the modulated signals transmitted by antenna  6202 A and by antenna  6202 B use at least frequency band X and frequency band Y. Frequency band X is used to transmit data of a first channel, and frequency band Y is used to transmit data of a second channel. 
     As described in Embodiment C1, the data of the first channel is transmitted using frequency band X in different data transmission mode. Accordingly, as shown in  FIG.  63   , the modulated signals transmitted by antenna  6202 A and by antenna  6202 B include components of frequency band X. These components of frequency band X are received by repeater A and by repeater B. Accordingly, as described in Embodiment 1 and in Embodiment C1, modulated signals in frequency band X are signals on which mapping has been performed, and to which precoding (weighting) and the change of phase are applied. 
     As shown in  FIG.  62   , the data of the second channel is transmitted by antenna  6202 A of  FIG.  2    and transmits data in components of frequency band Y. These components of frequency band Y are received by repeater A and by repeater B. 
       FIG.  64    illustrate the frequency allocation for transmit signals transmitted by repeater A and repeater B, specifically for modulated signal  6209 A transmitted by antenna  6210 A and modulated signal  6211 A transmitted by antenna  6212 A of repeater  6210 A, and for modulated signal  6209 B transmitted by antenna  6210 B and modulated signal  6211 B transmitted by antenna  6212 B of repeater B. In  FIG.  64   , frequency is on the horizontal axis and transmission power is on the vertical axis. 
     As shown, modulated signal  6209 A transmitted by antenna  6210 A and modulated signal  6211 A transmitted by antenna  6212 A use at least frequency band X and frequency band Y. Also, modulated signal  6209 B transmitted by antenna  6210 B and modulated signal  6211 B transmitted by antenna  6212 B similarly use at least frequency band X and frequency band Y. Frequency band X is used to transmit data of a first channel, and frequency band Y is used to transmit data of a second channel. 
     As described in Embodiment C1, the data of the first channel is transmitted using frequency band X in different data transmission mode. Accordingly, as shown in  FIG.  64   , modulated signal  6209 A transmitted by antenna  6210 A and modulated signal  6211 A transmitted by antenna  6212 B include components of frequency band X. These components of frequency band X are received by terminal P. Similarly, as shown in  FIG.  64   , modulated signal  6209 B transmitted by antenna  6210 B and modulated signal  6211 B transmitted by antenna  6212 B include components of frequency band X. These components of frequency band X are received by terminal R. Accordingly, as described in Embodiment 1 and in Embodiment C1, modulated signals in frequency band X are signals on which mapping has been performed, and to which precoding (weighting) and the change of phase are applied. 
     As shown in  FIG.  64   , the data of the second channel is carried by the modulated signals transmitted by antenna  6210 A of repeater A ( 6203 A) and by antenna  6210 B of repeater B ( 6203 ) from  FIG.  62    and transmits data in components of frequency band Y. Here, the components of frequency band Y in modulated signal  6209 A transmitted by antenna  6210 A of repeater A ( 6203 A) and those in modulated signal  6209 B transmitted by antenna  6210 B of repeater B ( 6203 B) are used in a transmission mode that involves identical data transmission, as explained in Embodiment C1. These components of frequency band Y are received by terminal Q. 
     The following describes the configuration of repeater A ( 6203 A) and repeater B ( 6203 B) from  FIG.  62   , with reference to  FIG.  65   . 
       FIG.  65    illustrates a sample configuration of a receiver and transmitter in a repeater. Components operating identically to those of  FIG.  56    use the same reference numbers thereas. Receiver  6203 X takes received signal  6502 A received by receive antenna  6501 A and received signal  6502 B received by receive antenna  6501 B as input, performs signal processing (signal demultiplexing or compositing, error-correction decoding, and so on) on the components of frequency band X thereof to obtain data  6204 X transmitted by the base station using frequency band X, outputs the data to the distributor  404  and obtains transmission method information included in control information (and transmission method information when transmitted by a repeater), and outputs the frame configuration signal  313 . 
     Receiver  6203 X and onward constitute a processor for generating a modulated signal for transmitting frequency band X. Further, the receiver here described is not only the receiver for frequency band X as shown in  FIG.  65   , but also incorporates receivers for other frequency bands. Each receiver forms a processor for generating modulated signals for transmitting a respective frequency band. 
     The overall operations of the distributor  404  are identical to those of the distributor in the base station described in Embodiment C2. 
     When transmitting as indicated in  FIG.  64   , repeater A ( 6203 A) and repeater B ( 6203 B) generate two different modulated signals (on which precoding and change of phase are performed) in frequency band X as described in Embodiment C1. The two modulated signals are respectively transmitted by antennas  6210 A and  6212 A of repeater A ( 6203 ) from  FIG.  62    and by antennas  6210 B and  6212 B of repeater B ( 6203 B) from  FIG.  62   . 
     As for frequency band Y, repeater A ( 6203 A) operates a processor  6500  pertaining to frequency band Y and corresponding to the signal processor  6500  pertaining to frequency band X shown in  FIG.  65    (the signal processor  6500  is the signal processor pertaining to frequency band X, but given that an identical signal processor is incorporated for frequency band Y, this description uses the same reference numbers), interleaver  304 A, mapper  306 A, weighting unit  308 A, and phase changer  5201  to generate modulated signal  5202 . A transmit signal corresponding to modulated signal  5202  is then transmitted by antenna  1301 A from  FIG.  13   , that is, by antenna  6210 A from  FIG.  62   . Similarly, repeater B ( 6203  B) operates interleaver  304 A, mapper  306 A, weighting unit  308 A, and phase changer  5201  from  FIG.  62    pertaining to frequency band Y to generate modulated signal  5202 . Then, a transmit signal corresponding to modulated signal  5202  is transmitted by antenna  1310 A from  FIG.  13   , i.e., by antenna  6210 B from  FIG.  62   . 
     As shown in  FIG.  66    ( FIG.  66    illustrates the frame configuration of the modulated signal transmitted by the base station, with time on the horizontal axis and frequency on the vertical axis), the base station transmits transmission method information  6601 , repeater-applied phase change information  6602 , and data symbols  6603 . The repeater obtains and applies the transmission method information  6601 , the repeater-applied phase change information  6602 , and the data symbols  6603  to the transmit signal, thus determining the phase changing method. When the repeater-applied phase change information  6602  from  FIG.  66    is not included in the signal transmitted by the base station, then as shown in  FIG.  62   , repeater B ( 6203 B) is the master and indicates the phase changing method to repeater A ( 6203 A). 
     As explained above, when the repeater transmits different data, the precoding matrix and phase changing method are set according to the transmission method to generate modulated signals. 
     On the other hand, to transmit identical data, two repeaters respectively generate and transmit modulated signals. In such circumstances, repeaters each generating modulated signals for transmission from a common antenna may be considered to be two combined repeaters using the precoding matrix given by Math. 52 (formula 52). The phase changing method is as explained in Embodiment C1, for example, and satisfies the conditions of Math. 53 (formula 53). 
     Also, as explained in Embodiment C1 for frequency band X, the base station and repeater may each have two antennas that transmit respective modulated signals and two antennas that receive identical data. The operations of such a base station or repeater are as described for Embodiment C1. 
     According to the present Embodiment, not only can the reception device obtain improved data reception quality for identical data transmission as well as different data transmission, but the transmission devices can also share a phase changer. 
     Furthermore, although the present Embodiment discusses examples using OFDM as the transmission method, the invention is not limited in this manner. Multi-carrier methods other than OFDM and single-carrier methods may all be used to achieve similar Embodiments. Here, spread-spectrum communications may also be used. When single-carrier methods are used, the change of phase is performed with respect to the time domain. 
     As explained in Embodiment 3, when the transmission method involves different data transmission, the change of phase is carried out on the data symbols, only. However, as described in the present Embodiment, when the transmission method involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.) 
     Embodiment C4 
     The present Embodiment concerns a phase changing method different from the phase changing methods described in Embodiment 1 and in the Supplement. 
     In Embodiment 1, Math. 36 (formula 36) is given as an example of a precoding matrix, and in the Supplement, Math. 50 (formula 50) is similarly given as another such example. In Embodiment A1, the phase changers from  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    are indicated as having a phase changing value of PHASE[i] (where i=0, 1, 2, . . . , N−2, N−1 (i being an integer between 0 and N−1)) to achieve a period (cycle) of N (value reached given that  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    perform a change of phase on only one baseband signal). The present description discusses performing a change of phase on one precoded baseband signal (i.e., in  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51  and  53   ) namely on precoded baseband signal z 2 ′. Here, PHASE[k] is calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     54 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       
                         k 
                         ⁢ 
                         π 
                       
                       N 
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     radians 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     54 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). 
     Accordingly, the reception device is able to achieve improvements in data reception quality in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the change of phase has not been performed, a regular phase relationship occurs. However, when the change of phase is performed, the phase relationship is modified, in turn avoiding poor conditions in a burst-like propagation environment. As an alternative to Math. 54 (formula 54), PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     55 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       - 
                       
                         
                           k 
                           ⁢ 
                           π 
                         
                         N 
                       
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     radians 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     55 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). 
     As a further alternative phase changing method, PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     56 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       
                         k 
                         ⁢ 
                         π 
                       
                       N 
                     
                     + 
                     
                       Z 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       radians 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     56 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). 
     As a further alternative phase changing method, PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     57 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       - 
                       
                         
                           k 
                           ⁢ 
                           π 
                         
                         N 
                       
                     
                     + 
                     
                       Z 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       radians 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     57 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). 
     As such, by performing the change of phase according to the present Embodiment, the reception device is made more likely to obtain good reception quality. 
     The change of phase of the present Embodiment is applicable not only to single-carrier methods but also to multi-carrier methods. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As previously described, while the present Embodiment explains the change of phase as a change of phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the change of phase with respect to the time domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing method of the present Embodiment is also applicable to a change of phase with respect to both the time domain and the frequency domain. Further, when the phase changing method described in the present Embodiment satisfies the conditions indicated in Embodiment A1, the reception device is highly likely to obtain good data quality. 
     Embodiment C5 
     The present Embodiment concerns a phase changing method different from the phase changing methods described in Embodiment 1, in the Supplement, and in Embodiment C4. 
     In Embodiment 1, Math. 36 (formula 36) is given as an example of a precoding matrix, and in the Supplement, Math. 50 (formula 50) is similarly given as another such example. In Embodiment A1, the phase changers from  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    are indicated as having a phase changing value of PHASE[i] (where i=0, 1, 2, . . . , N−2, N−1 (i being an integer between 0 and N−1)) to achieve a period (cycle) of N (value reached given that  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    perform a change of phase on only one baseband signal). The present description discusses performing a change of phase on one precoded baseband signal (i.e., in  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51  and  53   ) namely on precoded baseband signal z 2 ′. 
     The characteristic feature of the phase changing method pertaining to the present Embodiment is the period (cycle) of N=2π+1. To achieve the period (cycle) of N=2π+1, n+1 different phase changing values are prepared. Among these n+1 different phase changing values, n phase changing values are used twice per period (cycle), and one phase changing value is used only once per period (cycle), thus achieving the period (cycle) of N=2π+1. The following describes these phase changing values in detail. 
     The n+1 different phase changing values required to achieve a phase changing method in which the phase changing value is regularly switched in a period (cycle) of N=2π+1 are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[i] . . . PHASE[n−1], PHASE[n] (where i=0, 1, 2 . . . n−2, n−1, n (i being an integer between 0 and n)). Here, the n+1 different phase changing values of PHASE[ 0 ], PHASE[ 1 ], PHASE[i] . . . PHASE[n−1], PHASE[n] are expressed as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     58 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       
                         2 
                         ⁢ 
                         k 
                         ⁢ 
                         π 
                       
                       
                         
                           2 
                           ⁢ 
                           n 
                         
                         + 
                         1 
                       
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     radians 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     58 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , n−2, n−1, n (k being an integer between 0 and n). The n+1 different phase changing values PHASE[ 0 ], PHASE[ 1 ] . . . PHASE[i] PHASE[n−1], PHASE[n] are given by Math. 58 (formula 58). PHASE[ 0 ] is used once, while PHASE[ 1 ] through PHASE[n] are each used twice (i.e., PHASE[ 1 ] is used twice, PHASE[ 2 ] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing method in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing method is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are smaller, the effect thereof on the transmission device and reception device may be reduced. According to the above, the reception device is able to achieve improvements in data reception quality in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the change of phase has not been performed, a regular phase relationship occurs. However, when the change of phase is performed, the phase relationship is modified, in turn avoiding poor conditions in a burst-like propagation environment. As an alternative to Math. 58 (formula 58), PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     59 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       - 
                       
                         
                           2 
                           ⁢ 
                           k 
                           ⁢ 
                           π 
                         
                         
                           
                             2 
                             ⁢ 
                             n 
                           
                           + 
                           1 
                         
                       
                     
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     radians 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     5 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     9 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , n−2, n−1, n (k being an integer between 0 and n). 
     The n+1 different phase changing values PHASE[ 0 ], PHASE[ 1 ] PHASE[i] . . . PHASE[n−1], PHASE[n] are given by Math. 59 (formula 59). PHASE[ 0 ] is used once, while PHASE[ 1 ] through PHASE[n] are each used twice (i.e., PHASE[ 1 ] is used twice, PHASE[ 2 ] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing method in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing method is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are smaller, the effect thereof on the transmission device and reception device may be reduced. 
     As a further alternative, PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     60 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       
                         2 
                         ⁢ 
                         k 
                         ⁢ 
                         π 
                       
                       
                         
                           2 
                           ⁢ 
                           n 
                         
                         + 
                         1 
                       
                     
                     + 
                     
                       Z 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       radians 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     60 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , N−2, N−1 (k being an integer between 0 and N−1). 
     The n+1 different phase changing values PHASE[ 0 ], PHASE[ 1 ] . . . PHASE[i] . . . PHASE[n−1], PHASE[n] are given by Math. 60 (formula 60). PHASE[ 0 ] is used once, while PHASE[ 1 ] through PHASE[n] are each used twice (i.e., PHASE[ 1 ] is used twice, PHASE[ 2 ] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing method in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing method is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are smaller, the effect thereof on the transmission device and reception device may be reduced. 
     As a further alternative, PHASE[k] may be calculated as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     61 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     PHASE 
                     ⁢ 
                     
                         
                     
                     [ 
                     k 
                     ] 
                   
                   = 
                   
                     
                       - 
                       
                         
                           2 
                           ⁢ 
                           k 
                           ⁢ 
                           π 
                         
                         
                           
                             2 
                             ⁢ 
                             n 
                           
                           + 
                           1 
                         
                       
                     
                     + 
                     
                       Z 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       radians 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     61 
                   
                   ) 
                 
               
             
           
         
       
     
     where k=0, 1, 2, . . . , n−2, n−1, n (k being an integer between 0 and n). 
     The n+1 different phase changing values PHASE[ 0 ], PHASE[ 1 ] . . . PHASE[i] . . . PHASE[n−1], PHASE[n] are given by Math. 61 (formula 61). PHASE[ 0 ] is used once, while PHASE[ 1 ] through PHASE[n] are each used twice (i.e., PHASE[ 1 ] is used twice, PHASE[ 2 ] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing method in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing method is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are smaller, the effect thereof on the transmission device and reception device may be reduced. 
     As such, by performing the change of phase according to the present Embodiment, the reception device is made more likely to obtain good reception quality. 
     The change of phase of the present Embodiment is applicable not only to single-carrier methods but also to transmission using multi-carrier methods. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. As previously described, while the present Embodiment explains the change of phase as a change of phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the change of phase with respect to the time domain t described in the present Embodiment and replacing t with f (f being the ((sub-) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing method of the present Embodiment is also applicable to a change of phase with respect to both the time domain and the frequency domain. 
     Embodiment C6 
     The present Embodiment describes a method of regularly changing the phase, specifically that of Embodiment C5, when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC LDPC Codes (not only QC-LDPC but also LDPC codes may be used), concatenated LDPC (blocks) and BCH codes, Turbo codes or Duo-Binary Turbo codes using tail-biting, and so on. The following example considers a case where two streams s 1  and s 2  are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. 
       FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used.  FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  4   , and the transmission device has only one encoder. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  34   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     Then, given that the transmission device from  FIG.  4    transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s 1  and the other 1500 symbols are assigned to s 2 . As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s 1  and s 2 . 
     By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up each coded block, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up each coded block. 
     The following describes the relationship between the above-defined slots and the phase, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from  FIG.  4    uses five phase changing values (or phase changing sets) to achieve the period (cycle) of five. However, as described in Embodiment C5, three different phase changing values are present. Accordingly, some of the five phase changing values needed for the period (cycle) of five are identical. (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). The five phase changing values (or phase changing sets) needed for the period (cycle) of five are expressed as P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ]. 
     The following describes the relationship between the above-defined slots and the phase, as pertains to methods for a regular change of phase. 
     For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, phase changing value P[ 0 ] is used on 300 slots, phase changing value P[ 1 ] is used on 300 slots, phase changing value P[ 2 ] is used on 300 slots, phase changing value P[ 3 ] is used on 300 slots, and phase changing value P[ 4 ] is used on 300 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality. 
     Similarly, for the above-described 1500 slots needed to transmit the 6000 bits making up the pair of coded blocks when the modulation scheme is 16-QAM, phase changing value P[ 0 ] is used on 150 slots, phase changing value P[ 1 ] is used on 150 slots, phase changing value P[ 2 ] is used on 150 slots, phase changing value P[ 3 ] is used on 150 slots, and phase changing value P[ 4 ] is used on 150 slots. 
     Further, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, phase changing value P[ 0 ] is used on 100 slots, phase changing value P[ 1 ] is used on 100 slots, phase changing value P[ 2 ] is used on 100 slots, phase changing value P[ 3 ] is used on 100 slots, and phase changing value P[ 4 ] is used on 100 slots. 
     As described above, a phase changing method for regularly varying the phase changing value as given in Embodiment C5 requires the preparation of N=2n+1 phase changing values P[ 0 ], P[ 1 ] . . . P[ 2   n −1], P[ 2   n ] (where P[ 0 ], P[ 1 ] . . . P[ 2   n −1], P[ 2   n ] are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, in order to transmit all of the bits making up the coded block, phase changing value P[ 0 ] is used on K 0  slots, phase changing value P[ 1 ] is used on K 1  slots, phase changing value P[i] is used on Ki slots (where i=0, 1, 2, . . . , 2n−1, 2n (i being an integer between 0 and 2n)), and phase changing value P[ 2   n ] is used on K 2n  slots, such that Condition #C01 is met. 
     (Condition #C01) 
     K 0 =K 1  . . . =K i =K 2n . That is, K a =K b  (∀a and ∀b where a, b,=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     A phase changing method for a regular change of phase changing value as given in Embodiment C5 having a period (cycle) of N=2n+1 requires the preparation of phase changing values PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bits making up a single coded block, phase changing value PHASE[ 0 ] is used on G 0  slots, phase changing value PHASE[ 1 ] is used on G 1  slots, phase changing value PHASE[i] is used on G slots (where i=0, 1, 2, . . . , n−1, n (i being an integer between 0 and n)), and phase changing value PHASE[n] is used on G n  slots, such that Condition #C01 is met. Condition #C01 may be modified as follows. 
     (Condition #C02) 
     2×G 0 =G i = . . . G n . That is, 2×G 0 =G a  (∀a where α=1, 2 . . . n−1, n (a being an integer between 1 and n). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Condition #C01 (or Condition #C02) is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C01 (or Condition #C02) may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #C01. 
     (Condition #C03) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1, 2n(a and b being integers between 0 and 2n), a≠b). 
     Alternatively, Condition #C03 may be expressed as follows. 
     (Condition #C04) 
     The difference between G a  and G b  satisfies 0, 1, or 2. That is, |G a −G b | satisfies 0, 1, or 2(∀a, ∀b, where a, b=1, 2 . . . n−1, n (a and b being integers between 1 and n), a≠b) 
     and 
     The difference between 2×G 0  and G a  satisfies 0, 1, or 2. That is, |2×G 0 −G a | satisfies 0, 1, or 2(∀a, where a=1, 2 . . . n−1, n(a being an integer between 1 and n)). 
       FIG.  35    illustrates the varying numbers of symbols and slots needed in two coded blocks when block codes are used.  FIG.  35    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  3    and  FIG.  12   , and the transmission device has two encoders. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  35   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     The transmission device from  FIG.  3    and the transmission device from  FIG.  12    each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. Accordingly, when the modulation scheme is QPSK, two coded blocks drawn from s 1  and s 2  are transmitted within the same interval, e.g., a first coded block drawn from s 1  is transmitted, then a second coded block drawn from s 2  is transmitted. As such, 3000 slots are needed in order to transmit the first and second coded blocks. 
     By the same reasoning, when the modulation scheme is 16-QAM, 1500 slots are needed to transmit all of the bits making up two coded blocks, and when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks. 
     The following describes the relationship between the above-defined slots and the phase, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from  FIG.  4    uses five phase changing values (or phase changing sets) to achieve the period (cycle) of five. However, as described in Embodiment C5, three different phase changing values are present. Accordingly, some of the five phase changing values needed for the period (cycle) of five are identical. (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). The five phase changing values (or phase changing sets) needed for the period (cycle) of five are expressed as P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ]. 
     For the above-described 3000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is QPSK, phase changing value P[ 0 ] is used on 600 slots, phase changing value P[ 1 ] is used on 600 slots, phase changing value P[ 2 ] is used on 600 slots, phase changing value P[ 3 ] is used on 600 slots, and phase changing value P[ 4 ] is used on 600 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality. 
     Further, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 600 times, phase changing value P[ 1 ] is used on slots 600 times, phase changing value P[ 2 ] is used on slots 600 times, phase changing value P[ 3 ] is used on slots 600 times, and phase changing value PHASE[ 4 ] is used on slots 600 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 600 times, phase changing value P[ 1 ] is used on slots 600 times, phase changing value P[ 2 ] is used on slots 600 times, phase changing value P[ 3 ] is used on slots 600 times, and phase changing value P[ 4 ] is used on slots 600 times. 
     Similarly, for the above-described 1500 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 16-QAM, phase changing value P[ 0 ] is used on 300 slots, phase changing value P[ 1 ] is used on 300 slots, phase changing value P[ 2 ] is used on 300 slots, phase changing value P[ 3 ] is used on 300 slots, and phase changing value P[ 4 ] is used on 300 slots. 
     Furthermore, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 300 times, phase changing value P[ 1 ] is used on slots 300 times, phase changing value P[ 2 ] is used on slots 300 times, phase changing value P[ 3 ] is used on slots 300 times, and phase changing value P[ 4 ] is used on slots 300 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 300 times, phase changing value P[ 1 ] is used on slots 300 times, phase changing value P[ 2 ] is used on slots 300 times, phase changing value P[ 3 ] is used on slots 300 times, and phase changing value P[ 4 ] is used on slots 300 times. 
     Similarly, for the above-described 1000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 64-QAM, phase changing value P[ 0 ] is used on 200 slots, phase changing value P[ 1 ] is used on 200 slots, phase changing value P[ 2 ] is used on 200 slots, phase changing value P[ 3 ] is used on 200 slots, and phase changing value P[ 4 ] is used on 200 slots. 
     Furthermore, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 200 times, phase changing value P[ 1 ] is used on slots 200 times, phase changing value P[ 2 ] is used on slots 200 times, phase changing value P[ 3 ] is used on slots 200 times, and phase changing value P[ 4 ] is used on slots 200 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 200 times, phase changing value P[ 1 ] is used on slots 200 times, phase changing value P[ 2 ] is used on slots 200 times, phase changing value P[ 3 ] is used on slots 200 times, and phase changing value P[ 4 ] is used on slots 200 times. 
     As described above, a phase changing method for regularly varying the phase changing value as given in Embodiment C5 requires the preparation of N=2n+1 phase changing values P[ 0 ], P[ 1 ] . . . P[ 2   n −1], P[ 2   n ] (where P[ 0 ], P[ 1 ] . . . P[ 2   n −1], P[ 2   n ] are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, in order to transmit all of the bits making up the two coded blocks, phase changing value P[ 0 ] is used on K 0  slots, phase changing value P[ 1 ] is used on K 1  slots, phase changing value P[i] is used on K i  slots (where i=0,1,2 . . . 2n−1, 2n (i being an integer between 0 and 2π)), and phase changing value P[ 2   n ] is used on K2n slots. 
     (Condition #C05) 
     K 0 =K 1  . . . =K i = . . . K 2n . That is, K a =K b  (∀a and ∀b where a, b,=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2≠a≠b). 
     In order to transmit all of the bits making up the first coded block, phase changing value P[ 0 ] is used K 0,1  times, phase changing value P[ 1 ] is used K 1,1  times, phase changing value P[i] is used K i,1  (where i=0, 1, 2 . . . 2n−1, 2n (i being an integer between 0 and 2n)), and phase changing value P[ 2   n ] is used K 2n,1  times. 
     (Condition #C06) 
     K 0,1 =K 1,1  . . . =K i,1 = . . . K 2n,1 . That is, K a,1 =K b,1  (∀a and ∀b where a, b,=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     In order to transmit all of the bits making up the second coded block, phase changing value P[ 0 ] is used K 0,2  times, phase changing value P[ 1 ] is used K 1,2  times, phase changing value P[i] is used K i,2  (where i=0, 1, 2 . . . 2n−1, 2n (i being an integer between 0 and 2n)), and phase changing value P[ 2   n ] is used K 2n,2  times. 
     (Condition #C07) 
     K 0,2 =K 1,2  . . . =K i,2 = . . . K 2n,2 . That is, K a,2 =K b,2  (∀a and ∀b where a, b,=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     A phase changing method for regularly varying the phase changing value as given in Embodiment C5 having a period (cycle) of N=2n+1 requires the preparation of phase changing values PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bits making up the two coded blocks, phase changing value PHASE[ 0 ] is used on G 0  slots, phase changing value PHASE[ 1 ] is used on G 1  slots, phase changing value PHASE[i] is used on G slots (where i=0, 1, 2 . . . n−1, n (i being an integer between 0 and n)), and phase changing value PHASE[n] is used on G n  slots, such that Condition #C05 is met. 
     (Condition #C08) 
     2×G 0 =G 1  . . . =G i = . . . G n . That is, 2×G 0 =G a  (∀a where a=1, 2 . . . n−1, n (a being an integer between 1 and n)). 
     In order to transmit all of the bits making up the first coded block, phase changing value PHASE[ 0 ] is used G 0,1  times, phase changing value PHASE[ 1 ] is used G 1,1  times, phase changing value PHASE[i] is used G i,1  (where i=0, 1, 2 . . . n−1, n (i being an integer between 0 and n)), and phase changing value PHASE[n] is used G n,1  times. 
     (Condition #C09) 
     2×G 0,1 =G 1,1  . . . =G i,1 = . . . G n,1 . That is, 2×G 0,1 =G a,1  (∀a where α=1, 2 . . . n−1, n (a being an integer between 1 and n)). 
     In order to transmit all of the bits making up the second coded block, phase changing value PHASE[ 0 ] is used G 0,2  times, phase changing value PHASE[ 1 ] is used G 1,2  times, phase changing value PHASE[i] is used G i,2  (where i=0, 1, 2 . . . n−1, n (i being an integer between 0 and n)), and phase changing value PHASE[n] is used G n,1  times.
 
(Condition #C10)
 
2×G 0,2 =G 1,2  . . . =G i,2 = . . . G n,2 . That is, 2×G 0,2 =G a,2  (∀a where a=1, 2 . . . n−1, n (a being an integer between 1 and n)).
 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Condition #C05, Condition #C06, and Condition #C07 (or Condition #C08, Condition #C09, and Condition #C10) is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C05, Condition #C06, and Condition #C07 (or Condition #C08, Condition #C09, and Condition #C10) may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #C05, Condition #C06, and Condition #C07. 
     (Condition #C11) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     (Condition #C12) 
     The difference between K a,1  and K b,1  satisfies 0 or 1. That is, |K a,1 −K b,1 | satisfies 0 or 1(∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     (Condition #C13) 
     The difference between K a,2  and K b,2  satisfies 0 or 1. That is, |K a,2 −K b,2 | satisfies 0 or 1(∀a, ∀b, where a, b=0, 1, 2 . . . 2n−1, 2n (a and b being integers between 0 and 2n), a≠b). 
     Alternatively, Condition #C11, Condition #C12, and Condition #C13 may be expressed as follows. 
     (Condition #C14) 
     The difference between G a  and G b  satisfies 0, 1, or 2. That is, |G a −G b | satisfies 0, 1, or 2(∀a, ∀b, where a, b=1, 2 . . . n−1, n (a and b being integers between 1 and n), a≠b) 
     and 
     The difference between 2×G 0  and G a  satisfies 0, 1, or 2. That is, |2×G 0 −G a | satisfies 0, 1, or 2 (∀a, where a=1, 2 . . . n−1, n (a being an integer between 1 and n)). 
     (Condition #C15) 
     The difference between G a,1  and G b,1  satisfies 0, 1, or 2. That is, |G a,1 −G b,1 | satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . . . n−1, n (a and b being integers between 1 and n), a≠b) 
     and 
     The difference between 2×G 0,1  and G a,1  satisfies 0, 1, or 2. That is, |2×G 0,1 −G a,1 | satisfies 0, 1, or 2 (∀a, where a=1, 2 . . . n−1, n (a being an integer between 1 and n)) 
     (Condition #C16) 
     The difference between G a,2  and G b,2  satisfies 0, 1, or 2. That is, |G a,2 −G b,2 | satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2 . . . n−1, n (a and b being integers between 1 and n), a≠b) 
     and 
     The difference between 2×G 0,2  and G a,2  satisfies 0, 1, or 2. That is, |2×G 0,2 −G a,2 | satisfies 0, 1, or 2 (∀a, where a=1, 2 . . . n−1, n (a being an integer between 1 and n)) 
     As described above, bias among the phase changing values being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase changing values. As such, data reception quality can be improved for the reception device. 
     In the present Embodiment, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the method for a regular change of phase. As such, N phase changing values (or phase changing sets) P[ 0 ], P[ 1 ], P[ 2 ] . . . P[N−2], and P[N−1] are prepared. However, schemes exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) P[ 0 ], P[ 1 ], P[ 2 ] . . . P[N−2], and P[N−1] may also change the phases of blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing scheme with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the above-described conditions are satisfied, quality data reception improvements are realizable for the reception device. 
     Furthermore, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods using a fixed precoding matrix, space-time block coding methods, single-stream transmission, and methods using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission methods. 
     As described in Non-Patent Literature 3, spatial multiplexing MIMO methods involve transmitting signals s 1  and s 2 , which are mapped using a selected modulation scheme, on each of two different antennas. MIMO methods using a fixed precoding matrix involve performing precoding only (with no change in phase). Further, space-time block coding methods are described in Non-Patent Literature 9, 16, and 17. Single-stream transmission methods involve transmitting signal s 1 , mapped with a selected modulation scheme, from an antenna after performing predetermined processing. 
     Schemes using multi-carrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multi-carrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, space-time block coding schemes, single-stream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub-)carrier group are preferably used to realize the present Embodiment. 
     When a change of phase by, for example, a phase changing value for P[i] of X radians is performed on only one precoded baseband signal, the phase changers of  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    multiply precoded baseband signal z 2 ′ by e jX . Then, when a change of phase by, for example, a phase changing set for P[i] of X radians and Y radians is performed on both precoded baseband signals, the phase changers from  FIGS.  26 ,  27 ,  28 ,  52 , and  54    multiply precoded baseband signal z 2 ′ by e jX  and multiply precoded baseband signal z 1 ′ by e jY . 
     Embodiment C7 
     The present Embodiment describes a method of regularly changing the phase, specifically as done in Embodiment A1 and Embodiment C6, when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC LDPC Codes (not only QC-LDPC but also LDPC (block) codes may be used), concatenated LDPC and BCH codes, Turbo codes or Duo-Binary Turbo codes, and so on. The following example considers a case where two streams s 1  and s 2  are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. 
       FIG.  34    illustrates the varying numbers of symbols and slots needed in one coded block when block codes are used.  FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  4   , and the transmission device has only one encoder. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  34   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     Then, given that the transmission device from  FIG.  4    transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s 1  and the other 1500 symbols are assigned to s 2 . As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s 1  and s 2 . 
     By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up two coded blocks, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up the two coded blocks. 
     The following describes the relationship between the above-defined slots and the phase, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase, which has a period (cycle) of five. The phase changing values (or phase changing sets) prepared in order to regularly change the phase with a period (cycle) of five are P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ]. However, P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ] should include at least two different phase changing values (i.e., P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ] may include identical phase changing values). (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). 
     For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, phase changing value P[ 0 ] is used on 300 slots, phase changing value P[ 1 ] is used on 300 slots, phase changing value P[ 2 ] is used on 300 slots, phase changing value P[ 3 ] is used on 300 slots, and phase changing value P[ 4 ] is used on 300 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality. 
     Further, for the above-described 750 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16-QAM, phase changing value P[ 0 ] is used on 150 slots, phase changing value P[ 1 ] is used on 150 slots, phase changing value P[ 2 ] is used on 150 slots, phase changing value P[ 3 ] is used on 150 slots, and phase changing value P[ 4 ] is used on 150 slots. 
     Further, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, phase changing value P[ 0 ] is used on 100 slots, phase changing value P[ 1 ] is used on 100 slots, phase changing value P[ 2 ] is used on 100 slots, phase changing value P[ 3 ] is used on 100 slots, and phase changing value P[ 4 ] is used on 100 slots. 
     As described above, the phase changing values used in the phase changing method regularly switching between phase changing values with a period (cycle) of N are expressed as P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1]. However, P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1] should include at least two different phase changing values (i.e., P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1] may include identical phase changing values). In order to transmit all of the bits making up a single coded block, phase changing value P[ 0 ] is used on K 0  slots, phase changing value P[ 1 ] is used on K 1  slots, phase changing value P[i] is used on K i  slots (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and phase changing value P[N−1] is used on K N−1  slots, such that Condition #C17 is met. 
     (Condition #C17) 
     K 0 =K 1  . . . =K i = . . . K N−1 . That is, K a =K b  (∀a and ∀b where a, b, =0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Condition #C17 is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C17 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #C17. 
     (Condition #C18) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
       FIG.  35    illustrates the varying numbers of symbols and slots needed in two coded blocks when block codes are used.  FIG.  35    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  3    and  FIG.  12   , and the transmission device has two encoders. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  35   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     The transmission device from  FIG.  3    and the transmission device from  FIG.  12    each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. Accordingly, when the modulation scheme is QPSK, two coded blocks drawn from s 1  and s 2  are transmitted within the same interval, e.g., a first coded block drawn from s 1  is transmitted, then a second coded block drawn from s 2  is transmitted. As such, 3000 slots are needed in order to transmit the first and second coded blocks. 
     By the same reasoning, when the modulation scheme is 16-QAM, 1500 slots are needed to transmit all of the bits making up two coded blocks, and when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks. 
     The following describes the relationship between the above-defined slots and the phase, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from  FIG.  4    uses five phase changing values (or phase changing sets) P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ] to achieve the period (cycle) of five. However, P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ] should include at least two different phase changing values (i.e., P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ] may include identical phase changing values). (As in  FIG.  6   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on precoded baseband signal z 2 ′ only. Also, as in  FIG.  26   , two phase changing values are needed for each slot in order to perform the change of phase on both precoded baseband signals z 1 ′ and z 2 ′. These two phase changing values are termed a phase changing set. Accordingly, five phase changing sets should ideally be prepared in order to perform a change of phase having a period (cycle) of five in such circumstances). The five phase changing values (or phase changing sets) needed for the period (cycle) of five are expressed as P[ 0 ], P[ 1 ], P[ 2 ], P[ 3 ], and P[ 4 ]. 
     For the above-described 3000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is QPSK, phase changing value P[ 0 ] is used on 600 slots, phase changing value P[ 1 ] is used on 600 slots, phase changing value P[ 2 ] is used on 600 slots, phase changing value P[ 3 ] is used on 600 slots, and phase changing value P[ 4 ] is used on 600 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality. 
     Further, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 600 times, phase changing value P[ 1 ] is used on slots 600 times, phase changing value P[ 2 ] is used on slots 600 times, phase changing value P[ 3 ] is used on slots 600 times, and phase changing value PHASE[ 4 ] is used on slots 600 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 600 times, phase changing value P[ 1 ] is used on slots 600 times, phase changing value P[ 2 ] is used on slots 600 times, phase changing value P[ 3 ] is used on slots 600 times, and phase changing value P[ 4 ] is used on slots 600 times. 
     Similarly, for the above-described 1500 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 16-QAM, phase changing value P[ 0 ] is used on 300 slots, phase changing value P[ 1 ] is used on 300 slots, phase changing value P[ 2 ] is used on 300 slots, phase changing value P[ 3 ] is used on 300 slots, and phase changing value P[ 4 ] is used on 300 slots. 
     Furthermore, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 300 times, phase changing value P[ 1 ] is used on slots 300 times, phase changing value P[ 2 ] is used on slots 300 times, phase changing value P[ 3 ] is used on slots 300 times, and phase changing value P[ 4 ] is used on slots 300 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 300 times, phase changing value P[ 1 ] is used on slots 300 times, phase changing value P[ 2 ] is used on slots 300 times, phase changing value P[ 3 ] is used on slots 300 times, and phase changing value P[ 4 ] is used on slots 300 times. 
     Furthermore, for the above-described 1000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 64-QAM, phase changing value P[ 0 ] is used on 200 slots, phase changing value P[ 1 ] is used on 200 slots, phase changing value P[ 2 ] is used on 200 slots, phase changing value P[ 3 ] is used on 200 slots, and phase changing value P[ 4 ] is used on 200 slots. 
     Furthermore, in order to transmit the first coded block, phase changing value P[ 0 ] is used on slots 200 times, phase changing value P[ 1 ] is used on slots 200 times, phase changing value P[ 2 ] is used on slots 200 times, phase changing value P[ 3 ] is used on slots 200 times, and phase changing value P[ 4 ] is used on slots 200 times. Furthermore, in order to transmit the second coded block, phase changing value P[ 0 ] is used on slots 200 times, phase changing value P[ 1 ] is used on slots 200 times, phase changing value P[ 2 ] is used on slots 200 times, phase changing value P[ 3 ] is used on slots 200 times, and phase changing value P[ 4 ] is used on slots 200 times. 
     As described above, the phase changing values used in the phase changing method regularly switching between phase changing values with a period (cycle) of N are expressed as P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1]. However, P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1] should include at least two different phase changing values (i.e., P[ 0 ], P[ 1 ] . . . P[N−2], P[N−1] may include identical phase changing values). In order to transmit all of the bits making up a single coded block, phase changing value P[ 0 ] is used on K 0  slots, phase changing value P[ 1 ] is used on K 1  slots, phase changing value P[i] is used on K i  slots (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and phase changing value P[N−1] is used on K N−1  slots, such that Condition #C19 is met. 
     (Condition #C19) 
     K 0 =K 1  . . . =K i = . . . K N−1 . That is, K a =K b  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     In order to transmit all of the bits making up the first coded block, phase changing value P[ 0 ] is used K 0,1  times, phase changing value P[ 1 ] is used K 1 , 1 times, phase changing value P[i] is used K i,1  (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and phase changing value P[N−1] is used K N−1,1  times.
 
(Condition #C20)
 
K 0,1 =K 1,1 = . . . K i,1 = . . . K N−1,1 . That is, K a,1 =K b,1  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b).
 
     In order to transmit all of the bits making up the second coded block, phase changing value P[ 0 ] is used K 0,2  times, phase changing value P[ 1 ] is used K 1,2  times, phase changing value P[i] is used K i,2  (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and phase changing value P[N−1] is used K N−1,2  times. 
     (Condition #C21) 
     K 0,2 =K 1,2 = . . . K i,2 = . . . K N−1,2 . That is, K a,2 =K b,2  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1, a≠b). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Condition #C19, Condition #C20, and Condition #C21 are preferably met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C19, Condition #C20, and Condition #C21 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #C19, Condition #C20, and Condition #C21. 
     (Condition #C22) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     (Condition #C23) 
     The difference between K a,1  and K b,1  satisfies 0 or 1. That is, |K a,1 −K b,1 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b). 
     (Condition #C24) 
     The difference between K a,2  and K b,2  satisfies 0 or 1. That is, |K a,2 −K b,2 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1(a and b being integers between 0 and N−1), a≠b). 
     As described above, bias among the phase changing values being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase changing values. As such, data reception quality can be improved for the reception device. 
     In the present Embodiment, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the method for a regular change of phase. As such, N phase changing values (or phase changing sets) P[ 0 ], P[ 1 ], P[ 2 ] . . . P[N−2], and P[N−1] are prepared. However, methods exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) P[ 0 ], P[ 1 ], P[ 2 ] . . . P[N−2], and P[N−1] may also change the phases of blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing method with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the above-described conditions are satisfied, great quality data reception improvements are realizable for the reception device. 
     Furthermore, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods using a fixed precoding matrix, space-time block coding methods, single-stream transmission, and methods using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission methods. 
     As described in Non-Patent Literature 3, spatial multiplexing MIMO methods involve transmitting signals s 1  and s 2 , which are mapped using a selected modulation scheme, on each of two different antennas. MIMO methods using a fixed precoding matrix involve performing precoding only (with no change in phase). Further, space-time block coding methods are described in Non-Patent Literature 9, 16, and 17. Single-stream transmission methods involve transmitting signal s 1 , mapped with a selected modulation scheme, from an antenna after performing predetermined processing. 
     Schemes using multi-carrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multi-carrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, space-time block coding schemes, single-stream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub-)carrier group are preferably used to realize the present Embodiment. 
     When a change of phase by, for example, a phase changing value for P[i] of X radians is performed on only one precoded baseband signal, the phase changers of  FIGS.  3 ,  4 ,  6 ,  12 ,  25 ,  29 ,  51 , and  53    multiply precoded baseband signal z 2 ′ by e jX . Then, when a change of phase by, for example, a phase changing set for P[i] of X radians and Y radians is performed on both precoded baseband signals, the phase changers from  FIGS.  26 ,  27 ,  28 ,  52 , and  54    multiply precoded baseband signal z 2 ′ by e jX  and multiply precoded baseband signal z 1 ′ by e jY . 
     Embodiment D1 
     The present Embodiment is first described as a variation of Embodiment 1.  FIG.  67    illustrates a sample transmission device pertaining to the present Embodiment. Components thereof operating identically to those of  FIG.  3    use the same reference numbers thereas, and the description thereof is omitted for simplicity, below.  FIG.  67    differs from  FIG.  3    in the insertion of a baseband signal switcher  6702  directly following the weighting units. Accordingly, the following explanations are primarily centred on the baseband signal switcher  6702 . 
       FIG.  21    illustrates the configuration of the weighting units  308 A and  308 B. The area of  FIG.  21    enclosed in the dashed line represents one of the weighting units. Baseband signal  307 A is multiplied by w 11  to obtain w 11 ·s 1 ( t ), and multiplied by w 21  to obtain w 21 ·s 1 ( t ). Similarly, baseband signal  307 B is multiplied by w 12  to obtain w 12 ·s 2 ( t ), and multiplied by w 22  to obtain w 22 ·s 2 ( t ). Next, z 1 ( t )=w 11 ·s 1 ( t )+w 12 ·s 2 ( t ) and z 2 ( t )=w 21 ·s 1 ( t )+w 22 ·s 22 ( t ) are obtained. Here, as explained in Embodiment 1, s 1 ( t ) and s 2 ( t ) are baseband signals modulated according to a modulation scheme such as BPSK, QPSK, 8-PSK, 16-QAM, 32-QAM, 64-QAM, 256-QAM, 16-APSK and so on. Both weighting units perform weighting using a fixed precoding matrix. The precoding matrix uses, for example, the method of Math. 62 (formula 62), and satisfies the conditions of Math. 63 (formula 63) or Math. 64 (formula 64), all found below. However, this is only an example. The value of α is not limited to Math. 63 (formula 63) and Math. 64 (formula 64), and may, for example, be 1, or may be 0 (a is preferably a real number greater than or equal to 0, but may be also be an imaginary number). 
     Here, the precoding matrix is 
     
       
         
           
             
               
                 
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                       + 
                       
                         5 
                       
                     
                     
                       
                         2 
                       
                       + 
                       3 
                       - 
                       
                         5 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     64 
                   
                   ) 
                 
               
             
           
         
       
     
     Alternatively, the precoding matrix is not restricted to that of Math. 62 (formula 62), but may also be: 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     65 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     ( 
                     
                       
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             2 
                           
                         
                       
                       
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             2 
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     ( 
                     
                       
                         
                           a 
                         
                         
                           b 
                         
                       
                       
                         
                           c 
                         
                         
                           d 
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     65 
                   
                   ) 
                 
               
             
           
         
       
     
     where a=Ae jδ11 , b=Be jδ12 , c=Ce jδ21 , and d=De jδ22 . Further, one of a, b, c, and d may be equal to zero. For example: (1) a may be zero while b, c, and d are non-zero, (2) b may be zero while a, c, and d are non-zero, (3) c may be zero while a, b, and d are non-zero, or (4) d may be zero while a, b, and c are non-zero. 
     Alternatively, any two of a, b, c, and d may be equal to zero. For example, (1) a and d may be zero while b and c are non-zero, or (2) b and c may be zero while a and d are non-zero. 
     When any of the modulation scheme, error-correcting codes, and the coding rate thereof are changed, the precoding matrix in use may also be set and changed, or the same precoding matrix may be used as-is. 
     Next, the baseband signal switcher  6702  from  FIG.  67    is described. The baseband signal switcher  6702  takes weighted signal  309 A and weighted signal  316 B as input, performs baseband signal switching, and outputs switched baseband signal  6701 A and switched baseband signal  6701 B. The details of baseband signal switching are as described with reference to  FIG.  55   . The baseband signal switching performed in the present Embodiment differs from that of  FIG.  55    in terms of the signal used for switching. The following describes the baseband signal switching of the present Embodiment with reference to  FIG.  68   . 
     In  FIG.  68   , weighted signal  309 A(p 1 ( i )) has an in-phase component I of I p1 ( i ) and a quadrature component Q of Q p1 ( i ), while weighted signal  316 B(p 2 ( i )) has an in-phase component I of I p2 ( i ) and a quadrature component Q of Q p2 ( i ). In contrast, switched baseband signal  6701 A(q 1 ( i ) has an in-phase component I of I q1 ( i ) and a quadrature component Q of Q q1 ( i ), while switched baseband signal  6701 B(q 2 ( i ) has an in-phase component I of I q2 ( i ) and a quadrature component Q of Q q2 ( i ). (Here, i represents (time or (carrier) frequency order. In the example of  FIG.  67   , i represents time, though i may also represent (carrier) frequency when  FIG.  67    is applied to an OFDM scheme, as in  FIG.  12   . These points are elaborated upon below.) 
     Here, the baseband components are switched by the baseband signal switcher  6702 , such that:
         For switched baseband signal q 1 ( i ), the in-phase component I may be I p1 ( i ) while the quadrature component Q may be Q p2 ( i ), and for switched baseband signal q 2 ( i ), the in-phase component I may be I p2 ( i ) while the quadrature component q may be Q p1 ( i ). The modulated signal corresponding to switched baseband signal q 1 ( i ) is transmitted by transmit antenna  1  and the modulated signal corresponding to switched baseband signal q 2 ( i ) is transmitted from transmit antenna  2 , simultaneously on a common frequency. As such, the modulated signal corresponding to switched baseband signal q 1 ( i ) and the modulated signal corresponding to switched baseband signal q 2 ( i ) are transmitted from different antennas, simultaneously on a common frequency. Alternatively,   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be I p2  ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be Q p2  ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be Q p2  ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be I p2  ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be Q p2  ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be I p2  ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 1 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be I p2  ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be I p2  ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be Q p2  ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be Q p2  ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be I p2  ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be Q p2  ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 ( i ) while the quadrature component may be Q p2  ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be I p2 ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be I p2  ( i ) while the quadrature component may be Q p1 ( i ).   For switched baseband signal q 2 ( i ), the in-phase component may be Q p2  ( i ) while the quadrature component may be I p1 ( i ), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 ( i ) while the quadrature component may be I p2  ( i ).       

     Alternatively, the weighted signals  309 A and  316 B are not limited to the above-described switching of in-phase component and quadrature component. Switching may be performed on in-phase components and quadrature components greater than those of the two signals. 
     Also, while the above examples describe switching performed on baseband signals having a common timestamp (common (sub-)carrier) frequency), the baseband signals being switched need not necessarily have a common timestamp (common (sub-)carrier) frequency). For example, any of the following are possible.
         For switched baseband signal q 1 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be Q p2  (i+w), and for switched baseband signal q 2 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be I p2  (i+w), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be Q p2  (i+w).   For switched baseband signal q 1 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be Q p2  (i+w).   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be I p2  (i+w), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 1 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 1 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be Q p2  (i+w), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be I p2  (i+w).   For switched baseband signal q 1 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 2 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 1 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 2 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be I p2  (i+w).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be I p2  (i+w), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be Q p2  (i+w).   For switched baseband signal q 2 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be Q p2  (i+w).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be I p2  (i+w), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 2 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be Q p2  (i+w), and for switched baseband signal q 1 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 2 ( i ), the in-phase component may be I p1 (i+v) while the quadrature component may be Q p2  (i+w), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be I p2  (i+w).   For switched baseband signal q 2 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 1 ( i ), the in-phase component may be I p2  (i+w) while the quadrature component may be Q p1 (i+v).   For switched baseband signal q 2 ( i ), the in-phase component may be Q p2  (i+w) while the quadrature component may be I p1 (i+v), and for switched baseband signal q 1 ( i ), the in-phase component may be Q p1 (i+v) while the quadrature component may be I p2  (i+w).       

     Here, weighted signal  309 A(p 1 ( i )) has an in-phase component I of I p1 ( i ) and a quadrature component Q of Q p1 ( i ), while weighted signal  316 B(p 2 ( i )) has an in-phase component I of I p2  ( i ) and a quadrature component Q of Q p2  ( i ). In contrast, switched baseband signal  6701 A(q 1 ( i )) has an in-phase component I of I q1 (i) and a quadrature component Q of Q q1 (i), while switched baseband signal  6701 B(q 2 ( i )) has an in-phase component I q2 ( i ) and a quadrature component Q of Q q2 ( i ). 
     In  FIG.  68   , as described above, weighted signal  309 A(p 1 ( i )) has an in-phase component I of I p1 ( i ) and a quadrature component Q of Q p1 ( i ), while weighted signal  316 B(p 2 ( i )) has an in-phase component I of I p2  ( i ) and a quadrature component Q of Q p2  ( i ). In contrast, switched baseband signal  6701 A(q 1 ( i )) has an in-phase component I of LAO and a quadrature component Q of Q q1 ( i ), while switched baseband signal  6701 B(q 2 ( i )) has an in-phase component LAO and a quadrature component Q of Q q2 ( i ). 
     As such, in-phase component I of I q1 ( i ) and quadrature component Q of Q q1 ( i ) of switched baseband signal  6701 A(q 1 ( i )) and in-phase component I q2 ( i ) and quadrature component Q of Q q2 ( i ) of baseband signal  6701 B(q 2 ( i )) are expressible as any of the above. 
     As such, the modulated signal corresponding to switched baseband signal  6701 A(q 1 ( i )) is transmitted from transmit antenna  312 A, while the modulated signal corresponding to switched baseband signal  6701 B(q 2 ( i )) is transmitted from transmit antenna  312 B, both being transmitted simultaneously on a common frequency. Thus, the modulated signals corresponding to switched baseband signal  6701 A(q 1 ( i )) and switched baseband signal  6701 B(q 2 ( i )) are transmitted from different antennas, simultaneously on a common frequency. 
     Phase changer  317 B takes switched baseband signal  6701 B and signal processing method information  315  as input and regularly changes the phase of switched baseband signal  6701 B for output. This regular change is a change of phase performed according to a predetermined phase changing pattern having a predetermined period (cycle) (e.g., every n symbols (n being an integer, n≥1) or at a predetermined interval). The phase changing pattern is described in detail in Embodiment 4. 
     Wireless unit  310 B takes post-phase change signal  309 B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal  311 B. Transmit signal  311 B is then output as radio waves by an antenna  312 B. 
       FIG.  67   , much like  FIG.  3   , is described as having a plurality of encoders. However,  FIG.  67    may also have an encoder and a distributor like  FIG.  4   . In such a case, the signals output by the distributor are the respective input signals for the interleaver, while subsequent processing remains as described above for  FIG.  67   , despite the changes required thereby. 
       FIG.  5    illustrates an example of a frame configuration in the time domain for a transmission device according to the present Embodiment. Symbol  500 _ 1  is a symbol for notifying the reception device of the transmission method. For example, symbol  500 _ 1  conveys information such as the error-correction method used for transmitting data symbols, the coding rate thereof, and the modulation scheme used for transmitting data symbols. 
     Symbol  501 _ 1  is for estimating channel fluctuations for modulated signal z 1 ( t ) (where t is time) transmitted by the transmission device. Symbol  502 _ 1  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u (in the time domain). Symbol  503 _ 1  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u+1. 
     Symbol  501 _ 2  is for estimating channel fluctuations for modulated signal z 2 ( t ) (where t is time) transmitted by the transmission device. Symbol  502 _ 2  is a data symbol transmitted by modulated signal z 2 ( t ) as symbol number u. Symbol  503 _ 2  is a data symbol transmitted by modulated signal z 1 ( t ) as symbol number u+1. 
     Here, the symbols of z 1 ( t ) and of z 2 ( t ) having the same timestamp (identical timing) are transmitted from the transmit antenna using the same (shared/common) frequency. 
     The following describes the relationships between the modulated signals z 1 ( t ) and z 2 ( t ) transmitted by the transmission device and the received signals r 1 ( t ) and r 2 ( t ) received by the reception device. 
     In  FIGS.  5 ,  504    # 1  and  504  # 2  indicate transmit antennas of the transmission device, while  505  # 1  and  505  # 2  indicate receive antennas of the reception device. The transmission device transmits modulated signal z 1 ( t ) from transmit antenna  504  # 1  and transmits modulated signal z 2 ( t ) from transmit antenna  504  # 2 . Here, modulated signals z 1 ( t ) and z 2 ( t ) are assumed to occupy the same (shared/common) frequency (bandwidth). The channel fluctuations in the transmit antennas of the transmission device and the antennas of the reception device are h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ), respectively. Assuming that receive antenna  505  # 1  of the reception device receives received signal r 1 ( t ) and that receive antenna  505  # 2  of the reception device receives received signal r 2 ( t ), the following relationship holds. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     66 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     ( 
                     
                       
                         
                           
                             r 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             1 
                             ⁢ 
                             
                               ( 
                               t 
                               ) 
                             
                           
                         
                       
                       
                         
                           
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                             ⁢ 
                             2 
                             ⁢ 
                             
                               ( 
                               t 
                               ) 
                             
                           
                         
                       
                     
                     ) 
                   
                   = 
                   
                     
                       ( 
                       
                         
                           
                             
                               
                                 h 
                                 
                                   1 
                                   ⁢ 
                                   1 
                                 
                               
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                                 ( 
                                 t 
                                 ) 
                               
                             
                           
                           
                             
                               
                                 h 
                                 
                                   1 
                                   ⁢ 
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                                 ( 
                                 t 
                                 ) 
                               
                             
                           
                         
                         
                           
                             
                               
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                                   ⁢ 
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                                 ( 
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                                 ) 
                               
                             
                           
                           
                             
                               
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                                   ⁢ 
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                                 ( 
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                                 ) 
                               
                             
                           
                         
                       
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                     ⁢ 
                     
                       ( 
                       
                         
                           
                             
                               z 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
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                               ⁢ 
                               
                                 ( 
                                 t 
                                 ) 
                               
                             
                           
                         
                         
                           
                             
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                               ⁢ 
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                               ⁢ 
                               
                                 ( 
                                 t 
                                 ) 
                               
                             
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     66 
                   
                   ) 
                 
               
             
           
         
       
     
       FIG.  69    pertains to the weighting method (precoding method), the baseband switching method, and the phase changing method of the present Embodiment. The weighting unit  600  is a combined version of the weighting units  308 A and  308 B from  FIG.  67   . As shown, stream s 1 ( t ) and stream s 2 ( t ) correspond to the baseband signals  307 A and  307 B of  FIG.  3   . That is, the streams s 1 ( t ) and s 2 ( t ) are baseband signals made up of an in-phase component I and a quadrature component Q conforming to mapping by a modulation scheme such as QPSK, 16-QAM, and 64-QAM. As indicated by the frame configuration of  FIG.  69   , stream s 1 ( t ) is represented as s 1 ( u ) at symbol number u, as s 1 ( u +1) at symbol number u+1, and so forth. Similarly, stream s 2 ( t ) is represented as s 2 ( u ) at symbol number u, as s 2 ( u +1) at symbol number u+1, and so forth. The weighting unit  600  takes the baseband signals  307 A (s 1 ( t )) and  307 B (s 2 ( t )) as well as the signal processing method information  315  from  FIG.  67    as input, performs weighting in accordance with the signal processing method information  315 , and outputs the weighted signals  309 A ( p1 ( t )) and  316 B( p2 ( t )) from  FIG.  67   . 
     Here, given vector W 1 =(w 11 , w 12 ) from the first row of the fixed precoding matrix F, p 1 ( t ) can be expressed as Math. 67 (formula 67), below.
 
[Math. 67]
 
 p 1( t )= W 1 s 1( t )  (formula 67)
 
     Here, given vector W 2 =(w 21 ,w 22 ) from the first row of the fixed precoding matrix F, p 2 ( t ) can be expressed as Math. 68 (formula 68), below.
 
[Math. 68]
 
 p 2( t )= W 2 s 2( t )  (formula 68)
 
     Accordingly, precoding matrix F may be expressed as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     69 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   F 
                   = 
                   
                     ( 
                     
                       
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             1 
                           
                         
                         
                           
                             w 
                             ⁢ 
                             1 
                             ⁢ 
                             2 
                           
                         
                       
                       
                         
                           
                             w 
                             ⁢ 
                             2 
                             ⁢ 
                             1 
                           
                         
                         
                           
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                             ⁢ 
                             2 
                             ⁢ 
                             2 
                           
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     69 
                   
                   ) 
                 
               
             
           
         
       
     
     After the baseband signals have been switched, switched baseband signal  6701 A(q 1 ( i )) has an in-phase component I of Iq 1 ( i ) and a quadrature component Q of Qp 1 ( i ), and switched baseband signal  6701 B(q 2 ( i )) has an in-phase component I of Iq 2 ( i ) and a quadrature component Q of Qq 2 ( i ). The relationships between all of these are as stated above. When the phase changer uses phase changing formula y(t), the post-phase change baseband signal  309 B(q′ 2 ( i )) is given by Math. 70 (formula 70), below.
 
[Math. 70]
 
 q 2′( t )= y ( t ) q 2( t )  (formula 70)
 
     Here, y(t) is a phase changing formula obeying a predetermined method. For example, given a period (cycle) of four and timestamp u, the phase changing formula may be expressed as Math. 71 (formula 71), below.
 
[Math. 71]
 
 y ( u )= e   j0   (formula 71)
 
     Similarly, the phase changing formula for timestamp u+1 may be, for example, as given by Math. 72 (formula 72). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     72 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     y 
                     ⁡ 
                     
                       ( 
                       
                         u 
                         + 
                         1 
                       
                       ) 
                     
                   
                   = 
                   
                     e 
                     
                       j 
                       ⁢ 
                       
                         π 
                         2 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     72 
                   
                   ) 
                 
               
             
           
         
       
     
     That is, the phase changing formula for timestamp u+k generalizes to Math. 73 (formula 73). 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     73 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     y 
                     ⁡ 
                     
                       ( 
                       
                         u 
                         + 
                         k 
                       
                       ) 
                     
                   
                   = 
                   
                     e 
                     
                       j 
                       ⁢ 
                       
                         
                           k 
                           ⁢ 
                           π 
                         
                         2 
                       
                     
                   
                 
               
               
                 
                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     73 
                   
                   ) 
                 
               
             
           
         
       
     
     Note that Math. 71 (formula 71) through Math. 73 (formula 73) are given only as an example of a regular change of phase. 
     The regular change of phase is not restricted to a period (cycle) of four. Improved reception capabilities (the error-correction capabilities, to be exact) may potentially be promoted in the reception device by increasing the period (cycle) number (this does not mean that a greater period (cycle) is better, though avoiding small numbers such as two is likely ideal). 
     Furthermore, although Math. 71 (formula 71) through Math. 73 (formula 73), above, represent a configuration in which a change of phase is carried out through rotation by consecutive predetermined phases (in the above formula, every π/2), the change of phase need not be rotation by a constant amount but may also be random. For example, in accordance with the predetermined period (cycle) of y(t), the phase may be changed through sequential multiplication as shown in Math. 74 (formula 74) and Math. 75 (formula 75). The key point of the regular change of phase is that the phase of the modulated signal is regularly changed. The phase changing degree variance rate is preferably as even as possible, such as from −π radians to π radians. However, given that this concerns a distribution, random variance is also possible. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
                     . 
                     
                         
                     
                     ⁢ 
                     74 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       e 
                       
                         j 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         0 
                       
                     
                     → 
                     
                       e 
                       
                         j 
                         ⁢ 
                         
                           π 
                           5 
                         
                       
                     
                     → 
                     
                       e 
                       
                         j 
                         ⁢ 
                         
                           
                             2 
                             ⁢ 
                             π 
                           
                           5 
                         
                       
                     
                     → 
                     
                       e 
                       
                         j 
                         ⁢ 
                         
                           
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     As such, the weighting unit  600  of  FIG.  6    performs precoding using fixed, predetermined precoding weights, the baseband signal switcher performs baseband signal switching as described above, and the phase changer changes the phase of the signal input thereto while regularly varying the degree of change. 
     When a specialized precoding matrix is used in the LOS environment, the reception quality is likely to improve tremendously. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may greatly differ from the specialized precoding matrix, upon reception. The LOS environment has certain rules. Thus, data reception quality is tremendously improved through a regular change of transmit signal phase that obeys those rules. The present invention offers a signal processing method for improving the LOS environment. 
       FIG.  7    illustrates a sample configuration of a reception device  700  pertaining to the present embodiment. Wireless unit  703 _X receives, as input, received signal  702 _X received by antenna  701 _X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal  704 _X. 
     Channel fluctuation estimator  705 _ 1  for modulated signal z 1  transmitted by the transmission device takes baseband signal  704 _X as input, extracts reference symbol  501 _ 1  for channel estimation from  FIG.  5   , estimates the value of h 11  from Math. 66 (formula 66), and outputs channel estimation signal  706 _ 1 . 
     Channel fluctuation estimator  705 _ 2  for modulated signal z 2  transmitted by the transmission device takes baseband signal  704 _X as input, extracts reference symbol  501 _ 2  for channel estimation from  FIG.  5   , estimates the value of h 12  from Math. 66 (formula 66), and outputs channel estimation signal  706 _ 2 . 
     Wireless unit  703 _Y receives, as input, received signal  702 _Y received by antenna  701 _X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal  704 _Y. 
     Channel fluctuation estimator  707 _ 1  for modulated signal z 1  transmitted by the transmission device takes baseband signal  704 _Y as input, extracts reference symbol  501 _ 1  for channel estimation from  FIG.  5   , estimates the value of h 21  from Math. 66 (formula 66), and outputs channel estimation signal  708 _ 1 . 
     Channel fluctuation estimator  707 _ 2  for modulated signal z 2  transmitted by the transmission device takes baseband signal  704 _Y as input, extracts reference symbol  501 _ 2  for channel estimation from  FIG.  5   , estimates the value of h 22  from Math. 66 (formula 66), and outputs channel estimation signal  708 _ 2 . 
     A control information decoder  709  receives baseband signal  704 _X and baseband signal  704 _Y as input, detects symbol  500 _ 1  that indicates the transmission method from  FIG.  5   , and outputs a transmission device transmission method information signal  710 . 
     A signal processor  711  takes the baseband signals  704 _X and  704 _Y, the channel estimation signals  706 _ 1 ,  706 _ 2 ,  708 _ 1 , and  708 _ 2 , and the transmission method information signal  710  as input, performs detection and decoding, and then outputs received data  712 _ 1  and  712 _ 2 . 
     Next, the operations of the signal processor  711  from  FIG.  7    are described in detail.  FIG.  8    illustrates a sample configuration of the signal processor  711  pertaining to the present embodiment. As shown, the signal processor  711  is primarily made up of an inner MIMO detector, a soft-in/soft-out decoder, and a coefficient generator. Non-Patent Literature 2 and Non-Patent Literature 3 describe the method of iterative decoding with this structure. The MIMO system described in Non-Patent Literature 2 and Non-Patent Literature 3 is a spatial multiplexing MIMO system, while the present Embodiment differs from Non-Patent Literature 2 and Non-Patent Literature 3 in describing a MIMO system that regularly changes the phase over time, while using the precoding matrix and performing baseband signal switching. Taking the (channel) matrix H(t) of Math. 66 (formula 66), then by letting the precoding weight matrix from  FIG.  69    be F (here, a fixed precoding matrix remaining unchanged for a given received signal) and letting the phase changing formula used by the phase changer from  FIG.  69    be Y(t) (here, Y(t) changes over time t), then given the baseband signal switching, the receive vector R(t)=(r 1 ( t ),r 2 ( t )) T  and the stream vector S(t)=(s 1 ( t ),s 2 ( t )) T  lead to the decoding method of Non-Patent Literature 2 and Non-Patent Literature 3, thus enabling MIMO detection. 
     Accordingly, the coefficient generator  819  from  FIG.  8    takes a transmission method information signal  818  (corresponding to  710  from  FIG.  7   ) indicated by the transmission device (information for specifying the fixed precoding matrix in use and the phase changing pattern used when the phase is changed) and outputs a signal processing method information signal  820 . 
     The inner MIMO detector  803  takes the signal processing method information signal  820  as input and performs iterative detection and decoding using the signal. The operations are described below. 
     The processing unit illustrated in  FIG.  8    must use a processing method, as is illustrated in  FIG.  10   , to perform iterative decoding (iterative detection). First, detection of one codeword (or one frame) of modulated signal (stream) s 1  and of one codeword (or one frame) of modulated signal (stream) s 2  are performed. As a result, the soft-in/soft-out decoder obtains the log-likelihood ratio of each bit of the codeword (or frame) of modulated signal (stream) s 1  and of the codeword (or frame) of modulated signal (stream) s 2 . Next, the log-likelihood ratio is used to perform a second round of detection and decoding. These operations (referred to as iterative decoding (iterative detection)) are performed multiple times. The following explanations centre on the creation method of the log-likelihood ratio of a symbol at a specific time within one frame. 
     In  FIG.  8   , a memory  815  takes baseband signal  801 X (corresponding to baseband signal  704 _X from  FIG.  7   ), channel estimation signal group  802 X (corresponding to channel estimation signals  706 _ 1  and  706 _ 2  from  FIG.  7   ), baseband signal  801 Y (corresponding to baseband signal  704 _Y from  FIG.  7   ), and channel estimation signal group  802 Y (corresponding to channel estimation signals  708 _ 1  and  708 _ 2  from  FIG.  7   ) as input, performs iterative decoding (iterative detection), and stores the resulting matrix as a transformed channel signal group. The memory  815  then outputs the above-described signals as needed, specifically as baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, and transformed channel estimation signal group  817 Y. 
     Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection). 
     (Initial Detection) 
     The inner MIMO detector  803  takes baseband signal  801 X, channel estimation signal group  802 X, baseband signal  801 Y, and channel estimation signal group  802 Y as input. Here, the modulation scheme for modulated signal (stream) s 1  and modulated signal (stream) s 2  is described as 16-QAM. 
     The inner MIMO detector  803  first computes a candidate signal point corresponding to baseband signal  801 X from the channel estimation signal groups  802 X and  802 Y.  FIG.  11    represents such a calculation. In  FIG.  11   , each black dot is a candidate signal point in the I-Q plane. Given that the modulation scheme is 16-QAM, 256 candidate signal points exist. (However,  FIG.  11    is only a representation and does not indicate all 256 candidate signal points.) Letting the four bits transmitted in modulated signal s 1  be b 0 , b 1 , b 2 , and b 3  and the four bits transmitted in modulated signal s 2  be b 4 , b 5 , b 6 , and b 7 , candidate signal points corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) are found in  FIG.  11   . The Euclidean squared distance between each candidate signal point and each received signal point  1101  (corresponding to baseband signal  801 X) is then computed. The Euclidian squared distance between each point is divided by the noise variance σ 2 . Accordingly, E X (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is calculated. That is, the Euclidian squared distance between a candidate signal point corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) and a received signal point is divided by the noise variance. Here, each of the baseband signals and the modulated signals s 1  and s 2  is a complex signal. 
     Similarly, the inner MIMO detector  803  calculates candidate signal points corresponding to baseband signal  801 Y from channel estimation signal group  802 X and channel estimation signal group  802 Y, computes the Euclidean squared distance between each of the candidate signal points and the received signal points (corresponding to baseband signal  801 Y), and divides the Euclidean squared distance by the noise variance σ2. Accordingly, E Y (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is calculated. That is, E Y  is the Euclidian squared distance between a candidate signal point corresponding to (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) and a received signal point, divided by the noise variance. 
     Next, E X (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 )+E Y (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 )=E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is computed. 
     The inner MIMO detector  803  outputs E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) as the signal  804 . 
     Log-likelihood calculator  805 A takes the signal  804  as input, calculates the log-likelihood of bits b 0 , b 1 , b 2 , and b 3 , and outputs a log-likelihood signal  806 A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 and the log-likelihood of a bit being 0. The calculation method is as shown in Math. 28 (formula 28), Math. 29 (formula 29), and Math. 30 (formula 30), and the details are given by Non-Patent Literature 2 and 3. 
     Similarly, log-likelihood calculator  805 B takes the signal  804  as input, calculates the log-likelihood of bits b 4 , b 5 , b 6 , and b 7 , and outputs log-likelihood signal  806 B. 
     A deinterleaver ( 807 A) takes log-likelihood signal  806 A as input, performs deinterleaving corresponding to that of the interleaver (the interleaver ( 304 A) from  FIG.  67   ), and outputs deinterleaved log-likelihood signal  808 A. 
     Similarly, a deinterleaver ( 807 B) takes log-likelihood signal  806 B as input, performs deinterleaving corresponding to that of the interleaver (the interleaver ( 6704 B) from  FIG.  67   ), and outputs deinterleaved log-likelihood signal  808 B. 
     Log-likelihood ratio calculator  809 A takes deinterleaved log-likelihood signal  808 A as input, calculates the log-likelihood ratio of the bits encoded by encoder  6702 A from  FIG.  67   , and outputs log-likelihood ratio signal  810 A. 
     Similarly, log-likelihood ratio calculator  809 B takes deinterleaved log-likelihood signal  808 B as input, calculates the log-likelihood ratio of the bits encoded by encoder  302 B from  FIG.  67   , and outputs log-likelihood ratio signal  810 B. 
     Soft-in/soft-out decoder  811 A takes log-likelihood ratio signal  810 A as input, performs decoding, and outputs a decoded log-likelihood ratio  812 A. 
     Similarly, soft-in/soft-out decoder  811 B takes log-likelihood ratio signal  810 B as input, performs decoding, and outputs decoded log-likelihood ratio  812 B. 
     (Iterative Decoding (Iterative Detection), k Iterations) 
     The interleaver ( 813 A) takes the k−1th decoded log-likelihood ratio  812 A decoded by the soft-in/soft-out decoder as input, performs interleaving, and outputs interleaved log-likelihood ratio  814 A. Here, the interleaving pattern used by the interleaver ( 813 A) is identical to that of the interleaver ( 304 A) from  FIG.  67   . 
     Another interleaver ( 813 B) takes the k−1th decoded log-likelihood ratio  812 B decoded by the soft-in/soft-out decoder as input, performs interleaving, and outputs interleaved log-likelihood ratio  814 B. Here, the interleaving pattern used by the interleaver ( 813 B) is identical to that of the other interleaver ( 304 B) from  FIG.  67   . 
     The inner MIMO detector  803  takes baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, transformed channel estimation signal group  817 Y, interleaved log-likelihood ratio  814 A, and interleaved log-likelihood ratio  814 B as input. Here, baseband signal  816 X, transformed channel estimation signal group  817 X, baseband signal  816 Y, and transformed channel estimation signal group  817 Y are used instead of baseband signal  801 X, channel estimation signal group  802 X, baseband signal  801 Y, and channel estimation signal group  802 Y because the latter cause delays due to the iterative decoding. 
     The iterative decoding operations of the inner MIMO detector  803  differ from the initial detection operations thereof in that the interleaved log-likelihood ratios  814 A and  814 B are used in signal processing for the former. The inner MIMO detector  803  first calculates E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) in the same manner as for initial detection. In addition, the coefficients corresponding to Math. 11 (formula 11) and Math. 32 (formula 32) are computed from the interleaved log-likelihood ratios  814 A and  914 B. The value of E(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ) is corrected using the coefficients so calculated to obtain E′(b 0 , b 1 , b 2 , b 3 , b 4 , b 5 , b 6 , b 7 ), which is output as the signal  804 . 
     The log-likelihood calculator  805 A takes the signal  804  as input, calculates the log-likelihood of bits b 0 , b 1 , b 2 , and b 3 , and outputs the log-likelihood signal  806 A. Note that this log-likelihood calculation produces the log-likelihood of a bit being 1 and the log-likelihood of a bit being 0. The calculation method is as shown in Math. 31 (formula 31) through Math. 35 (formula 35), and the details are given by Non-Patent Literature 2 and 3. 
     Similarly, log-likelihood calculator  805 B takes the signal  804  as input, calculates the log-likelihood of bits b 4 , b 5 , b 6 , and b 7 , and outputs log-likelihood signal  806 B. Operations performed by the deinterleaver onwards are similar to those performed for initial detection. 
     While  FIG.  8    illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary as good reception improvements are obtainable by iterative detection alone. As long as the components needed for iterative detection are present, the configuration need not include the interleavers  813 A and  813 B. In such a case, the inner MIMO detector  803  does not perform iterative detection. 
     As shown in Non-Patent Literature 5 and the like, QR decomposition may also be used to perform initial detection and iterative detection. Also, as indicated by Non-Patent Literature 11, MMSE and ZF linear operations may be performed when performing initial detection. 
       FIG.  9    illustrates the configuration of a signal processor unlike that of  FIG.  8   , that serves as the signal processor for modulated signals transmitted by the transmission device from  FIG.  4    as used in  FIG.  67   . The point of difference from  FIG.  8    is the number of soft-in/soft-out decoders. A soft-in/soft-out decoder  901  takes the log-likelihood ratio signals  810 A and  810 B as input, performs decoding, and outputs a decoded log-likelihood ratio  902 . A distributor  903  takes the decoded log-likelihood ratio  902  as input for distribution. Otherwise, the operations are identical to those explained for  FIG.  8   . 
     As described above, when a transmission device according to the present Embodiment using a MIMO system transmits a plurality of modulated signals from a plurality of antennas, changing the phase over time while multiplying by the precoding matrix so as to regularly change the phase results in improvements to data reception quality for a reception device in a LOS environment, where direct waves are dominant, compared to a conventional spatial multiplexing MIMO system. 
     In the present Embodiment, and particularly in the configuration of the reception device, the number of antennas is limited and explanations are given accordingly. However, the Embodiment may also be applied to a greater number of antennas. In other words, the number of antennas in the reception device does not affect the operations or advantageous effects of the present Embodiment. 
     Further, in the present Embodiments, the encoding is not particularly limited to LDPC codes. Similarly, the decoding method is not limited to implementation by a soft-in/soft-out decoder using sum-product decoding. The decoding method used by the soft-in/soft-out decoder may also be, for example, the BCJR algorithm, S OVA, and the Max-Log-Map algorithm. Details are provided in Non-Patent Literature 6. 
     In addition, although the present Embodiment is described using a single-carrier method, no limitation is intended in this regard. The present Embodiment is also applicable to multi-carrier transmission. Accordingly, the present Embodiment may also be realized using, for example, spread-spectrum communications, OFDM, SC-FDMA, SC-OFDM, wavelet OFDM as described in Non-Patent Literature 7, and so on. Furthermore, in the present Embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, and so on) or symbols transmitting control information, may be arranged within the frame in any manner. 
     The following describes an example in which OFDM is used as a multi-carrier method. 
       FIG.  70    illustrates the configuration of a transmission device using OFDM. In  FIG.  70   , components operating in the manner described for  FIGS.  3 ,  12 , and  67    use identical reference numbers. 
     An OFDM-related processor  1201 A takes weighted signal  309 A as input, performs OFDM-related processing thereon, and outputs transmit signal  1202 A. Similarly, OFDM-related processor  1201 B takes post-phase change signal  309 B as input, performs OFDM-related processing thereon, and outputs transmit signal  1202 B. 
       FIG.  13    illustrates a sample configuration of the OFDM-related processors  1201 A and  1201 B and onward from  FIG.  70   . Components  1301 A through  1310 A belong between  1201 A and  312 A from  FIG.  70   , while components  1301 B through  1310 B belong between  1201 B and  312 B. 
     Serial-to-parallel converter  1302 A performs serial-to-parallel conversion on switched baseband signal  1301 A (corresponding to switched baseband signal  6701 A from  FIG.  70   ) and outputs parallel signal  1303 A. 
     Reorderer  1304 A takes parallel signal  1303 A as input, performs reordering thereof, and outputs reordered signal  1305 A. Reordering is described in detail later. 
     IFFT unit  1306 A takes reordered signal  1305 A as input, applies an IFFT thereto, and outputs post-IFFT signal  1307 A. 
     Wireless unit  1308 A takes post-IFFT signal  1307 A as input, performs processing such as frequency conversion and amplification, thereon, and outputs modulated signal  1309 A. Modulated signal  1309 A is then output as radio waves by antenna  1310 A. 
     Serial-to-parallel converter  1302 B performs serial-to-parallel conversion on post-phase change  1301 B (corresponding to post-phase change  309 B from  FIG.  12   ) and outputs parallel signal  1303 B. 
     Reorderer  1304 B takes parallel signal  1303 B as input, performs reordering thereof, and outputs reordered signal  1305 B. Reordering is described in detail later. 
     IFFT unit  1306 B takes reordered signal  1305 B as input, applies an IFFT thereto, and outputs post-IFFT signal  1307 B. 
     Wireless unit  1308 B takes post-IFFT signal  1307 B as input, performs processing such as frequency conversion and amplification thereon, and outputs modulated signal  1309 B. Modulated signal  1309 B is then output as radio waves by antenna  1310 A. 
     The transmission device from  FIG.  67    does not use a multi-carrier transmission method. Thus, as shown in  FIG.  69   , a change of phase is performed to achieve a period (cycle) of four and the post-phase change symbols are arranged in the time domain. As shown in  FIG.  70   , when multi-carrier transmission, such as OFDM, is used, then, naturally, symbols in precoded baseband signals having undergone switching and phase changing may be arranged in the time domain as in  FIG.  67   , and this may be applied to each (sub-)carrier. However, for multi-carrier transmission, the arrangement may also be in the frequency domain, or in both the frequency domain and the time domain. The following describes these arrangements. 
       FIGS.  14 A and  14 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13   . The frequency axes are made up of (sub-)carriers  0  through  9 . The modulated signals z 1  and z 2  share common timestamps (timing) and use a common frequency band.  FIG.  14 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  14 B  illustrates a reordering method for the symbols of modulated signal z 2 . With respect to the symbols of switched baseband signal  1301 A input to serial-to-parallel converter  1302 A, the ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, #0, #1, #2, and #3 are equivalent to one period (cycle). Similarly, #4n, #4n+1, #4n+2, and #4n+3 (n being a non-zero positive integer) are also equivalent to one period (cycle). 
     As shown in  FIG.  14 A , symbols #0, #1, #2, #3, and so on are arranged in order, beginning at carrier  0 . Symbols #0 through #9 are given timestamp $ 1 , followed by symbols #10 through #19 which are given timestamp #2, and so on in a regular arrangement. Here, modulated signals z 1  and z 2  are complex signals. 
     Similarly, with respect to the symbols of weighted signal  1301 B input to serial-to-parallel converter  1302 B, the assigned ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, a different change in phase is applied to each of #0, #1, #2, and #3, which are equivalent to one period (cycle). Similarly, a different change in phase is applied to each of #4n, #4n+1, #4n+2, and #4n+3 (n being a non-zero positive integer), which are also equivalent to one period (cycle). 
     As shown in  FIG.  14 B , symbols #0, #1, #2, #3, and so on are arranged in order, beginning at carrier  0 . Symbols #0 through #9 are given timestamp $ 1 , followed by symbols #10 through #19 which are given timestamp $ 2 , and so on in a regular arrangement. 
     The symbol group  1402  shown in  FIG.  14 B  corresponds to one period (cycle) of symbols when the phase changing method of  FIG.  69    is used. Symbol #0 is the symbol obtained by using the phase at timestamp u in  FIG.  69   , symbol #1 is the symbol obtained by using the phase at timestamp u+1 in  FIG.  69   , symbol #2 is the symbol obtained by using the phase at timestamp u+2 in  FIG.  69   , and symbol #3 is the symbol obtained by using the phase at timestamp u+3 in  FIG.  69   . Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at timestamp u in  FIG.  69    when x mod 4 equals 0 (i.e., when the remainder of x divided by 4 is 0, mod being the modulo operator), symbol #x is the symbol obtained by using the phase at timestamp x+1 in  FIG.  69    when x mod 4 equals 1, symbol #x is the symbol obtained by using the phase at timestamp x+2 in  FIG.  69    when x mod 4 equals 2, and symbol #x is the symbol obtained by using the phase at timestamp x+3 in  FIG.  69    when x mod 4 equals 3. 
     In the present Embodiment, modulated signal z 1  shown in  FIG.  14 A  has not undergone a change of phase. 
     As such, when using a multi-carrier transmission method such as OFDM, and unlike single carrier transmission, symbols can be arranged in the frequency domain. Of course, the symbol arrangement method is not limited to those illustrated by  FIGS.  14 A and  14 B . Further examples are shown in  FIGS.  15 A,  15 B,  16 A, and  16 B . 
       FIGS.  15 A and  15 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  14 A and  14 B .  FIG.  15 A  illustrates a reordering scheme for the symbols of modulated signal z 1 , while  FIG.  15 B  illustrates a reordering scheme for the symbols of modulated signal z 2 .  FIGS.  15 A and  15 B  differ from  FIGS.  14 A and  14 B  in that different reordering methods are applied to the symbols of modulated signal z 1  and to the symbols of modulated signal z 2 . In  FIG.  15 B , symbols #0 through #5 are arranged at carriers  4  through  9 , symbols #6 though #9 are arranged at carriers  0  through  3 , and this arrangement is repeated for symbols #10 through #19. Here, as in  FIG.  14 B , symbol group  1502  shown in  FIG.  15 B  corresponds to one period (cycle) of symbols when the phase changing method of  FIG.  6    is used. 
       FIGS.  16 A and  16 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  14 A and  14 B .  FIG.  16 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  16 B  illustrates a reordering method for the symbols of modulated signal z 2 .  FIGS.  16 A and  16 B  differ from  FIGS.  14 A and  14 B  in that, while  FIGS.  14 A and  14 B  showed symbols arranged at sequential carriers,  FIGS.  16 A and  16 B  do not arrange the symbols at sequential carriers. Obviously, for  FIGS.  16 A and  16 B , different reordering methods may be applied to the symbols of modulated signal z 1  and to the symbols of modulated signal z 2  as in  FIGS.  15 A and  15 B . 
       FIGS.  17 A and  17 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from those of  FIGS.  14 A through  16 B .  FIG.  17 A  illustrates a reordering method for the symbols of modulated signal z 1  and  FIG.  17 B  illustrates a reordering method for the symbols of modulated signal z 2 . While  FIGS.  14 A through  16 B  show symbols arranged with respect to the frequency axis,  FIGS.  17 A and  17 B  use the frequency and time axes together in a single arrangement. 
     While  FIG.  69    describes an example where the change of phase is performed in a four slot period (cycle), the following example describes an eight slot period (cycle). In  FIGS.  17 A and  17 B , the symbol group  1702  is equivalent to one period (cycle) of symbols when the phase changing scheme is used (i.e., to eight symbols) such that symbol #0 is the symbol obtained by using the phase at timestamp u, symbol #1 is the symbol obtained by using the phase at timestamp u+1, symbol #2 is the symbol obtained by using the phase at timestamp u+2, symbol #3 is the symbol obtained by using the phase at timestamp u+3, symbol #4 is the symbol obtained by using the phase at timestamp u+4, symbol #5 is the symbol obtained by using the phase at timestamp u+5, symbol #6 is the symbol obtained by using the phase at timestamp u+6, and symbol #7 is the symbol obtained by using the phase at timestamp u+7. Accordingly, for any symbol #x, symbol #x is the symbol obtained by using the phase at timestamp u when x mod 8 equals 0, symbol #x is the symbol obtained by using the phase at timestamp u+1 when x mod 8 equals 1, symbol #x is the symbol obtained by using the phase at timestamp u+2 when x mod 8 equals 2, symbol #x is the symbol obtained by using the phase at timestamp u+3 when x mod 8 equals 3, symbol #x is the symbol obtained by using the phase at timestamp u+4 when x mod 8 equals 4, symbol #x is the symbol obtained by using the phase at timestamp u+5 when x mod 8 equals 5, symbol #x is the symbol obtained by using the phase at timestamp u+6 when x mod 8 equals 6, and symbol #x is the symbol obtained by using the phase at timestamp u+7 when x mod 8 equals 7. In  FIGS.  17 A and  17 B  four slots along the time axis and two slots along the frequency axis are used for a total of 4×2=8 slots, in which one period (cycle) of symbols is arranged. Here, given m×n symbols per period (cycle) (i.e., m×n different phases are available for multiplication), then n slots (carriers) in the frequency domain and m slots in the time domain should be used to arrange the symbols of each period (cycle), such that m&gt;n. This is because the phase of direct waves fluctuates slowly in the time domain relative to the frequency domain. Accordingly, the present Embodiment performs a regular change of phase that reduces the influence of steady direct waves. Thus, the phase changing period (cycle) should preferably reduce direct wave fluctuations. Accordingly, m should be greater than n. Taking the above into consideration, using the time and frequency domains together for reordering, as shown in  FIGS.  17 A and  17 B , is preferable to using either of the frequency domain or the time domain alone due to the strong probability of the direct waves becoming regular. As a result, the effects of the present invention are more easily obtained. However, reordering in the frequency domain may lead to diversity gain due the fact that frequency-domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal. 
       FIGS.  18 A and  18 B  indicate frequency on the horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from that of  FIGS.  17 A and  17 B .  FIG.  18 A  illustrates a reordering method for the symbols of modulated signal z 1 , while  FIG.  18 B  illustrates a reordering method for the symbols of modulated signal z 2 . Much like  FIGS.  17 A and  17 B ,  FIGS.  18 A and  18 B  illustrate the use of the time and frequency domains, together. However, in contrast to  FIGS.  17 A and  17 B , where the frequency domain is prioritized and the time domain is used for secondary symbol arrangement,  FIGS.  18 A and  18 B  prioritize the time domain and use the frequency domain for secondary symbol arrangement. In  FIG.  18 B , symbol group  1802  corresponds to one period (cycle) of symbols when the phase changing method is used. 
     In  FIGS.  17 A,  17 B,  18 A, and  18 B , the reordering method applied to the symbols of modulated signal z 1  and the symbols of modulated signal z 2  may be identical or may differ as like in  FIGS.  15 A and  15 B . Either approach allows good reception quality to be obtained. Also, in  FIGS.  17 A,  17 B,  18 A, and  18 B , the symbols may be arranged non-sequentially as in  FIGS.  16 A and  16 B . Either approach allows good reception quality to be obtained. 
       FIG.  22    indicates frequency on the horizontal axis and time on the vertical axis thereof, and illustrates an example of a symbol reordering method used by the reorderers  1301 A and  1301 B from  FIG.  13    that differs from the above.  FIG.  22    illustrates a regular phase changing method using four slots, similar to timestamps u through u+3 from  FIG.  69   . The characteristic feature of  FIG.  22    is that, although the symbols are reordered with respect to the frequency domain, when read along the time axis, a periodic shift of n (n=1 in the example of  FIG.  22   ) symbols is apparent. The frequency-domain symbol group  2210  in  FIG.  22    indicates four symbols to which are applied the changes of phase at timestamps u through u+3 from  FIG.  69   . 
     Here, symbol #0 is obtained through a change of phase at timestamp u, symbol #1 is obtained through a change of phase at timestamp u+1, symbol #2 is obtained through a change of phase at timestamp u+2, and symbol #3 is obtained through a change of phase at timestamp u+3. 
     Similarly, for frequency-domain symbol group  2220 , symbol #4 is obtained through a change of phase at timestamp u, symbol #5 is obtained through a change of phase at timestamp u+1, symbol #6 is obtained through a change of phase at timestamp u+2, and symbol #7 is obtained through a change of phase at timestamp u+3. 
     The above-described change of phase is applied to the symbol at timestamp $ 1 . However, in order to apply periodic shifting with respect to the time domain, the following change of phases are applied to symbol groups  2201 ,  2202 ,  2203 , and  2204 . 
     For time-domain symbol group  2201 , symbol #0 is obtained through a change of phase at timestamp u, symbol #9 is obtained through a change of phase at timestamp u+1, symbol #18 is obtained through a change of phase at timestamp u+2, and symbol #27 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2202 , symbol #28 is obtained through a change of phase at timestamp u, symbol #1 is obtained through a change of phase at timestamp u+1, symbol #10 is obtained through a change of phase at timestamp u+2, and symbol #19 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2203 , symbol #20 is obtained through a change of phase at timestamp u, symbol #29 is obtained through a change of phase at timestamp u+1, symbol #2 is obtained through a change of phase at timestamp u+2, and symbol #11 is obtained through a change of phase at timestamp u+3. 
     For time-domain symbol group  2204 , symbol #12 is obtained through a change of phase at timestamp u, symbol #21 is obtained through a change of phase at timestamp u+1, symbol #30 is obtained through a change of phase at timestamp u+2, and symbol #3 is obtained through a change of phase at timestamp u+3. 
     The characteristic feature of  FIG.  22    is seen in that, taking symbol #11 as an example, the two neighbouring symbols thereof having the same timestamp in the frequency domain (#10 and #12) are both symbols changed using a different phase than symbol #11, and the two neighbouring symbols thereof having the same carrier in the time domain (#2 and #20) are both symbols changed using a different phase than symbol #11. This holds not only for symbol #11, but also for any symbol having two neighbouring symbols in the frequency domain and the time domain. Accordingly, the change of phase is effectively carried out. This is highly likely to improve data reception quality as influence from regularizing direct waves is less prone to reception. 
     Although  FIG.  22    illustrates an example in which n=1, the invention is not limited in this manner. The same may be applied to a case in which n=3. Furthermore, although  FIG.  22    illustrates the realization of the above-described effects by arranging the symbols in the frequency domain and advancing in the time domain so as to achieve the characteristic effect of imparting a periodic shift to the symbol arrangement order, the symbols may also be randomly (or regularly) arranged to the same effect. 
     Although the present Embodiment describes a variation of Embodiment 1 in which a baseband signal switcher is inserted before the change of phase, the present Embodiment may also be realized as a combination with Embodiment 2, such that the baseband signal switcher is inserted before the change of phase in  FIGS.  26  and  28   . Accordingly, in  FIG.  26   , phase changer  317 A takes switched baseband signal  6701 A(q 1 ( i )) as input, and phase changer  317 B takes switched baseband signal  6701 B(q 2 ( i )) as input. The same applies to the phase changers  317 A and  317 B from  FIG.  28   . 
     The following describes a method of allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device. 
       FIG.  31    illustrates an example of frame configuration for a portion of the symbols within a signal in the time-frequency domains, given a transmission method where a regular change of phase is performed for a multi-carrier method such as OFDM. 
       FIG.  31    illustrates the frame configuration of modulated signal z 2 ′ corresponding to the switched baseband signal input to phase changer  317 B from  FIG.  67   . Each square represents one symbol (although both signals s 1  and s 2  are included for precoding purposes, depending on the precoding matrix, only one of signals s 1  and s 2  may be used). 
     Consider symbol  3100  at carrier  2  and timestamp $ 2  of  FIG.  31   . The carrier here described may alternatively be termed a sub-carrier. 
     Within carrier  2 , there is a very strong correlation between the channel conditions for symbol  3100 A at carrier  2 , timestamp $ 2  and the channel conditions for the time domain nearest-neighbour symbols to timestamp $ 2 , i.e., symbol  3013  at timestamp $ 1  and symbol  3101  at timestamp $ 3  within carrier  2 . 
     Similarly, for timestamp $ 2 , there is a very strong correlation between the channel conditions for symbol  3100  at carrier  2 , timestamp $ 2  and the channel conditions for the frequency-domain nearest-neighbour symbols to carrier  2 , i.e., symbol  3104  at carrier  1 , timestamp $ 2  and symbol  3104  at timestamp $ 2 , carrier  3 . 
     As described above, there is a very strong correlation between the channel conditions for symbol  3100  and the channel conditions for each symbol  3101 ,  3102 ,  3103 , and  3104 . 
     The present description considers N different phases (N being an integer, N≥2) for multiplication in a transmission method where the phase is regularly changed. The symbols illustrated in  FIG.  31    are indicated as e j0 , for example. This signifies that this symbol is signal z 2 ′ from  FIG.  6    having undergone a change in phase through multiplication by e j0 . That is, the values given for the symbols in  FIG.  31    are the value of y(t) as given by Math. 70 (formula 70). 
     The present Embodiment takes advantage of the high correlation in channel conditions existing between neighbouring symbols in the frequency domain and/or neighbouring symbols in the time domain in a symbol arrangement enabling high data reception quality to be obtained by the reception device receiving the post-phase change symbols. 
     In order to achieve this high data reception quality, conditions #D1-1 and #D1-2 are met. 
     (Condition #D1-1) 
     As shown in  FIG.  69   , for a transmission method involving a regular change of phase performed on switched baseband signal q 2  using a multi-carrier method such as OFDM, time X, carrier Y is a symbol for transmitting data (hereinafter, data symbol), neighbouring symbols in the time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Y are also data symbols, and a different change of phase should be performed on switched baseband signal q 2  corresponding to each of these three data symbols, i.e., on switched baseband signal q 2  at time X, carrier Y, at time X−1, carrier Y and at time X+1, carrier Y.
 
(Condition #D1-2)
 
As shown in  FIG.  69   , for a transmission method involving a regular change of phase performed on switched baseband signal q 2  using a multi-carrier method such as OFDM, time X, carrier Y is a symbol for transmitting data (hereinafter, data symbol), neighbouring symbols in the time domain, i.e., at time X, carrier Y+1 and at time X, carrier Y−1 are also data symbols, and a different change of phase should be performed on switched baseband signal q 2  corresponding to each of these three data symbols, i.e., on switched baseband signal q 2  at time X, carrier Y, at time X, carrier Y−1 and at time X, carrier Y+1.
 
     Ideally, a data symbol should satisfy Condition #D1-1. Similarly, the data symbols should satisfy Condition #D1-2. 
     The reasons supporting Conditions #D1-1 and #D1-2 are as follows. 
     A very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above. 
     Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to phase relations despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding. 
     Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (symbol A) and the channel conditions of the symbols neighbouring symbol A in the frequency domain, as described above. 
     Accordingly, when three neighbouring symbols in the frequency domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding. 
     By combining Conditions #D1-1 and #D1-2, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition #D1-3 can be derived. 
     (Condition #D1-3) 
     As shown in  FIG.  69   , for a transmission method involving a regular change of phase performed on switched baseband signal q 2  using a multi-carrier method such as OFDM, time X, carrier Y is a symbol for transmitting data (data symbol), neighbouring symbols in the time domain, i.e., at time X−1, carrier Y and at time X+1, carrier Y are also data symbols, and neighbouring symbols in the frequency domain, i.e., at time X, carrier Y−1 and at time X, carrier Y+1 are also data symbols, such that a different change of phase should be performed on switched baseband signal q 2  corresponding to each of these five data symbols, i.e., on switched baseband signal q 2  at time X, carrier Y, at time X, carrier Y−1, at time X, carrier Y+1, at time X−1, carrier Y and at time X+1, carrier Y. 
     Here, the different changes in phase are as follows. Phase changes are defined from 0 radians to 2π radians. For example, for time X, carrier Y, a phase change of e jθX,Y  is applied to precoded baseband signal q 2  from  FIG.  69   , for time X−1, carrier Y, a phase change of e jθX−1,Y  is applied to precoded baseband signal q 2  from  FIG.  69   , for time X+1, carrier Y, a phase change of e jθX+1,Y  is applied to precoded baseband signal q 2  from  FIG.  69   , such that 0≤θ X,Y &lt;2π, 0≤θ X−1,Y ≤2π, and 0≤θ X+1,Y &lt;2π, all units being in radians. Accordingly, for Condition #D1-1, it follows that θ X,Y ≠θ X,Y−1 , θ X,Y ≠θ X,Y+1 , and that θ X,Y−1 ≠θ X,Y+1 . Similarly, for Condition #D1-2, it follows that θ X,Y ≠θ X−1,Y , θ X,Y ≠θ X+1 , and that θ X,Y−1 ≠θ X,Y+1 . And, for Condition #D1-3, it follows that θ X,Y ≠θ X−1,Y , θ X,Y ≠θ X+1,Y , θ X,Y ≠θ X,Y−1 , θ X,Y ≠θ X,Y+1 , θ X−1,Y ≠θ X+1,Y , θ X−1,Y ≠θ X−1,Y ·θ X,Y+1 , θ X+1,Y ≠θ X,Y−1 , θ X+1,Y ≠θ X,Y+1 , and that θ X,Y−1 ≠θ X,Y+1 . 
     Ideally, a data symbol should satisfy Condition #D1-3. 
       FIG.  31    illustrates an example of Condition #D1-3, where symbol A corresponds to symbol  3100 . The symbols are arranged such that the phase by which switched baseband signal q 2  from  FIG.  69    is multiplied differs for symbol  3100 , for both neighbouring symbols thereof in the time domain  3101  and  3102 , and for both neighbouring symbols thereof in the frequency domain  3102  and  3104 . Accordingly, despite received signal quality degradation of symbol  3100  for the receiver, good signal quality is highly likely for the neighbouring signals, thus guaranteeing good signal quality after error correction. 
       FIG.  32    illustrates a symbol arrangement obtained through phase changes under these conditions. 
     As evident from  FIG.  32   , with respect to any data symbol, a different change in phase is applied to each neighbouring symbol in the time domain and in the frequency domain. As such, the ability of the reception device to correct errors may be improved. 
     In other words, in  FIG.  32   , when all neighbouring symbols in the time domain are data symbols, Condition #D1-1 is satisfied for all Xs and all Ys. 
     Similarly, in  FIG.  32   , when all neighbouring symbols in the frequency domain are data symbols, Condition #D1-2 is satisfied for all Xs and all Ys. 
     Similarly, in  FIG.  32   , when all neighbouring symbols in the frequency domain are data symbols and all neighbouring symbols in the time domain are data symbols, Condition #D1-3 is satisfied for all Xs and all Ys. 
     The following discusses the above-described example for a case where the change of phase is performed on two switched baseband signals q 1  and q 2  (see  FIG.  68   ). 
     Several phase changing methods are applicable to performing a change of phase on two switched baseband signals q 1  and q 2 . The details thereof are explained below. 
     Scheme 1 involves a change in phase of switched baseband signal q 2  as described above, to achieve the change in phase illustrated by  FIG.  32   . In  FIG.  32   , a change of phase having a period (cycle) of ten is applied to switched baseband signal q 2 . However, as described above, in order to satisfy Conditions #D1-1, #D1-2, and #D1-3, the change in phase applied to switched baseband signal q 2  at each (sub-)carrier changes over time. (Although such changes are applied in  FIG.  32    with a period (cycle) of ten, other phase changing methods are also applicable.) Then, as shown in  FIG.  33   , the phase change degree performed on switched baseband signal q 2  produce a constant value that is one-tenth that of the change in phase performed on switched baseband signal q 2 . In  FIG.  33   , for a period (cycle) (of phase change performed on switched baseband signal q 2 ) including timestamp $ 1 , the value of the change in phase performed on switched baseband signal q 1  is e j0 . Then, for the next period (cycle) (of change in phase performed on switched baseband signal q 2 ) including timestamp $ 2 , the value of the phase changing degree performed on precoded baseband signal q 1  is e jπ/9 , and so on. 
     The symbols illustrated in  FIG.  33    are indicated as e j0 , for example. This signifies that this symbol is signal q 1  from  FIG.  26    having undergone a change of phase through multiplication by e j0 . 
     As shown in  FIG.  33   , the change in phase applied to switched baseband signal q 1  produces a constant value that is one-tenth that of the change in phase performed on precoded, switched baseband signal q 2  such that the post-phase change value varies with the number of each period (cycle). (As described above, in  FIG.  33   , the value is e j0  for the first period (cycle), e jπ/9  for the second period (cycle), and so on.) 
     As described above, the change in phase performed on switched baseband signal q 2  has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the degree of phase change applied to switched baseband signal q 1  and to switched baseband signal q 2  into consideration. Accordingly, data reception quality may be improved for the reception device. 
     Scheme 2 involves a change in phase of switched baseband signal q 2  as described above, to achieve the change in phase illustrated by  FIG.  32   . In  FIG.  32   , a change of phase having a period (cycle) of ten is applied to switched baseband signal q 2 . However, as described above, in order to satisfy Conditions #D1-1, #D1-2, and #D1-3, the change in phase applied to switched baseband signal q 2  at each (sub-)carrier changes over time. (Although such changes are applied in  FIG.  32    with a period (cycle) of ten, other phase changing methods are also applicable.) Then, as shown in  FIG.  33   , the change in phase performed on switched baseband signal q 2  produces a constant value that is one-tenth of that performed on switched baseband signal q 2 . 
     The symbols illustrated in  FIG.  30    are indicated as e j0 , for example. This signifies that this symbol is switched baseband signal q 1  having undergone a change of phase through multiplication by e j0 . 
     As described above, the change in phase performed on switched baseband signal q 2  has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the changes in phase applied to switched baseband signal q 1  and to switched baseband signal q 2  into consideration. Accordingly, data reception quality may be improved for the reception device. An effective way of applying method 2 is to perform a change in phase on switched baseband signal q 1  with a period (cycle) of N and perform a change in phase on precoded baseband signal q 2  with a period (cycle) of M such that N and M are coprime. As such, by taking both switched baseband signals q 1  and q 2  into consideration, a period (cycle) of N×M is easily achievable, effectively making the period (cycle) greater when N and M are coprime. 
     While the above discusses an example of the above-described phase changing method, the present invention is not limited in this manner. The change in phase may be performed with respect to the frequency domain, the time domain, or on time-frequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases. 
     The same also applies to frames having a configuration other than that described above, where pilot symbols (SP symbols) and symbols transmitting control information are inserted among the data symbols. The details of the change in phase in such circumstances are as follows. 
       FIGS.  47 A and  47 B  illustrate the frame configuration of modulated signals (switched baseband signals q 1  and q 2 ) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  47 A  illustrates the frame configuration of modulated signal (switched baseband signal q 1 ) z 1  or z 1 ′ while  FIG.  47 B  illustrates the frame configuration of modulated signal (switched baseband signal q 2 ) z 2 ′. In  FIGS.  47 A and  47 B,  4701    marks pilot symbols while  4702  marks data symbols. The data symbols  4702  are symbols on which switching or switching and change in phase have been performed. 
       FIGS.  47 A and  47 B , like  FIG.  69   , indicate the arrangement of symbols when a change in phase is applied to switched baseband signal q 2  (while no change in phase is performed on switched baseband signal q 1 ). (Although  FIG.  69    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  69    corresponds to a change in phase with respect to the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  47 A and  47 B  for each of the symbols are the values of switched baseband signal q 2  after the change in phase. No values are given for the symbols of switched baseband signal q 1  (z 1 ) from  FIGS.  47 A and  47 B  as no change in phase is performed thereon. 
     The important point of  FIGS.  47 A and  47 B  is that the change in phase performed on the data symbols of switched baseband signal q 2 , i.e., on symbols having undergone precoding or precoding and switching. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 2 ′. 
       FIGS.  48 A and  48 B  illustrate the frame configuration of modulated signals (switched baseband signals q 1  and q 2 ) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  48 A  illustrates the frame configuration of modulated signal (switched baseband signal q 1 ) z 1  or z 1 ′ while  FIG.  48 B  illustrates the frame configuration of modulated signal (switched baseband signal q 2 ) z 2 ′. In  FIGS.  48 A and  48 B,  4701    marks pilot symbols while  4702  marks data symbols. The data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed. 
       FIGS.  48 A and  48 B  indicate the arrangement of symbols when a change in phase is applied to switched baseband signal q 1  and to switched baseband signal q 2 . Accordingly, the numerical values indicated in  FIGS.  48 A and  48 B  for each of the symbols are the values of switched baseband signals q 1  and q 2  after a change in phase. 
     The important point of  FIGS.  48 A and  48 B  is that the change in phase is performed on the data symbols of switched baseband signal q 1 , that is, on the precoded or precoded and switched symbols thereof, and on the data symbols of switched baseband signal q 2 , that is, on the precoded or precoded and switched symbols thereof. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 1 ′, nor on the pilot symbols inserted in z 2 ′. 
       FIGS.  49 A and  49 B  illustrate the frame configuration of modulated signals (switched baseband signals q 1  and q 2 ) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  49 A  illustrates the frame configuration of modulated signal (switched baseband signal q 1 ) z 1  or z 1 ′ while  FIG.  49 B  illustrates the frame configuration of modulated signal (switched baseband signal q 2 ) z 2 ′. In  FIGS.  49 A and  49 B,  4701    marks pilot symbols,  4702  marks data symbols, and  4901  marks null symbols for which the in-phase component of the baseband signal I=0 and the quadrature component Q=0. As such, data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed.  FIGS.  49 A and  49 B  differ from  FIGS.  47 A and  47 B  in the configuration scheme for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal z 1 ′ are null symbols in modulated signal z 2 ′. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal z 2 ′ are null symbols in modulated signal z 1 ′. 
       FIGS.  49 A and  49 B , like  FIG.  69   , indicate the arrangement of symbols when a change in phase is applied to switched baseband signal q 2  (while no change in phase is performed on switched baseband signal q 1 ). (Although  FIG.  69    illustrates a change in phase with respect to the time domain, switching time t with carrier f in  FIG.  6    corresponds to a change in phase with respect to the frequency domain. In other words, replacing ( t ) with (t, f) where t is time and f is frequency corresponds to performing a change of phase on time-frequency blocks.) Accordingly, the numerical values indicated in  FIGS.  49 A and  49 B  for each of the symbols are the values of switched baseband signal q 2  after the change in phase. No values are given for the symbols of switched baseband signal q 1  from  FIGS.  49 A and  49 B  as no change in phase is performed thereon. 
     The important point of  FIGS.  49 A and  49 B  is that the change in phase performed on the data symbols of switched baseband signal q 2 , i.e., on symbols having undergone precoding or precoding and switching. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 2 ′. 
       FIGS.  50 A and  50 B  illustrate the frame configuration of modulated signals (switched baseband signals q 1  and q 2 ) z 1  or z 1 ′ and z 2 ′ in the time-frequency domain.  FIG.  50 A  illustrates the frame configuration of modulated signal (switched baseband signal q 1 ) z 1  or z 1 ′ while  FIG.  50 B  illustrates the frame configuration of modulated signal (switched baseband signal q 2 ) z 2 ′. In  FIGS.  50 A and  50 B,  4701    marks pilot symbols,  4702  marks data symbols, and  4901  marks null symbols for which the in-phase component of the baseband signal I=0 and the quadrature component Q=0. As such, data symbols  4702  are symbols on which precoding or precoding and a change in phase have been performed.  FIGS.  50 A and  50 B  differ from  FIGS.  48 A and  48 B  in the configuration scheme for symbols other than data symbols. The times and carriers at which pilot symbols are inserted into modulated signal z 1 ′ are null symbols in modulated signal z 2 ′. Conversely, the times and carriers at which pilot symbols are inserted into modulated signal z 2 ′ are null symbols in modulated signal z 1 ′. 
       FIGS.  50 A and  50 B  indicate the arrangement of symbols when a change in phase is applied to switched baseband signal q 1  and to switched baseband signal q 2 . Accordingly, the numerical values indicated in  FIGS.  50 A and  50 B  for each of the symbols are the values of switched baseband signals q 1  and q 2  after a change in phase. 
     The important point of  FIGS.  50 A and  50 B  is that a change in phase is performed on the data symbols of switched baseband signal q 1 , that is, on the precoded or precoded and switched symbols thereof, and on the data symbols of switched baseband signal q 2 , that is, on the precoded or precoded and switched symbols thereof. (The symbols under discussion, being precoded, actually include both symbols s 1  and s 2 .) Accordingly, no change in phase is performed on the pilot symbols inserted in z 1 ′, nor on the pilot symbols inserted in z 2 ′. 
       FIG.  51    illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the frame configuration of  FIGS.  47 A,  47 B,  49 A, and  49 B . Components thereof performing the same operations as those of  FIG.  4    use the same reference symbols thereas.  FIG.  51    does not include a baseband signal switcher as illustrated in  FIGS.  67  and  70   . However,  FIG.  51    may also include a baseband signal switcher between the weighting unit and phase changer, much like  FIGS.  67  and  70   . 
     In  FIG.  51   , the weighting units  308 A and  308 B, phase changer  317 B, and baseband signal switcher only operate at times indicated by the frame configuration signal  313  as corresponding to data symbols. 
     In  FIG.  51   , a pilot symbol generator  5101  (that also generates null symbols) outputs baseband signals  5102 A and  5102 B for a pilot symbol whenever the frame configuration signal  313  indicates a pilot symbol (and a null symbol). 
     Although not indicated in the frame configurations from  FIGS.  47 A through  50 B , when precoding (and phase rotation) is not performed, such as when transmitting a modulated signal using only one antenna (such that the other antenna transmits no signal) or when using a space-time coding transmission method (particularly, space-time block coding) to transmit control information symbols, then the frame configuration signal  313  takes control information symbols  5104  and control information  5103  as input. When the frame configuration signal  313  indicates a control information symbol, baseband signals  5102 A and  5102 B thereof are output. 
     Wireless units  310 A and  310 B of  FIG.  51    take a plurality of baseband signals as input and select a desired baseband signal according to the frame configuration signal  313 . The wireless units  310 A and  310 B then apply OFDM signal processing and output modulated signals  311 A and  311 B conforming to the frame configuration. 
       FIG.  52    illustrates a sample configuration of a transmission device generating and transmitting modulated signal having the frame configuration of  FIGS.  48 A,  48 B,  50 A, and  50 B . Components thereof performing the same operations as those of  FIGS.  4  and  51    use the same reference symbols thereas.  FIG.  52    features an additional phase changer  317 A that only operates when the frame configuration signal  313  indicates a data symbol. At all other times, the operations are identical to those explained for  FIG.  51   .  FIG.  52    does not include a baseband signal switcher as illustrated in  FIGS.  67  and  70   . However,  FIG.  52    may also include a baseband signal switcher between the weighting unit and phase changer, much like  FIGS.  67  and  70   . 
       FIG.  53    illustrates a sample configuration of a transmission device that differs from that of  FIG.  51   .  FIG.  53    does not include a baseband signal switcher as illustrated in  FIGS.  67  and  70   . However,  FIG.  53    may also include a baseband signal switcher between the weighting unit and phase changer, much like  FIGS.  67  and  70   . The following describes the points of difference. As shown in  FIG.  53   , phase changer  317 B takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  317 B performs the change in phase on precoded baseband signal  316 B. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  317 B pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     A selector  5301  takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal  313  for output. 
       FIG.  54    illustrates a sample configuration of a transmission device that differs from that of  FIG.  52   .  FIG.  54    does not include a baseband signal switcher as illustrated in  FIGS.  67  and  70   . However,  FIG.  54    may also include a baseband signal switcher between the weighting unit and phase changer, much like  FIGS.  67  and  70   . The following describes the points of difference. As shown in  FIG.  54   , phase changer  317 B takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  317 B performs the change in phase on precoded baseband signal  316 B. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  317 B pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     Similarly, as shown in  FIG.  54   , phase changer  5201  takes a plurality of baseband signals as input. Then, when the frame configuration signal  313  indicates a data symbol, phase changer  5201  performs the change in phase on precoded baseband signal  309 A. When frame configuration signal  313  indicates a pilot symbol (or null symbol) or a control information symbol, phase changer  5201  pauses phase changing operations such that the symbols of the baseband signal are output as-is. (This may be interpreted as performing forced rotation corresponding to e j0 .) 
     The above explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in this manner. When symbols are transmitted using methods other than precoding, such as single-antenna transmission or transmission using space-time block coding, the absence of change in phase is important. Conversely, performing the change of phase on symbols that have been precoded is the key point of the present invention. 
     Accordingly, a characteristic feature of the present invention is that the change in phase is not performed on all symbols within the frame configuration in the time-frequency domain, but only performed on baseband signals that have been precoded and have undergone switching. 
     The following describes a scheme for regularly changing the phase when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC LDPC Codes (not only QC-LDPC but also LDPC codes may be used), concatenated LDPC and BCH codes, Turbo codes or Duo-Binary Turbo codes using tail-biting, and so on. The following example considers a case where two streams s 1  and s 2  are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. 
       FIG.  34    illustrates the varying numbers of symbols and slots needed in two coded blocks when block codes are used. Unlike  FIGS.  69  and  70   , for example,  FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated in  FIG.  4   , with an encoder and distributor. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  34   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     Then, given that the above-described transmission device transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s 1  and the other 1500 symbols are assigned to s 2 . As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s 1  and s 2 . 
     By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up two coded blocks, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up the two coded blocks. 
     The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase. That is, the phase changer of the above-described transmission device uses five phase changing values (or phase changing sets) to achieve the period (cycle) of five. (As in  FIG.  69   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on switched baseband signal q 2  only. Similarly, in order to perform the change in phase on both switched baseband signals q 1  and q 2 , two phase changing values are needed for each slot. These two phase changing values are termed a phase changing set. Accordingly, here, in order to perform a change of phase having a period (cycle) of five, five such phase changing sets should be prepared). The five phase changing values (or phase changing sets) are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ], PHASE[ 3 ], and PHASE [ 4 ]. 
     For the above-described 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[ 0 ] is used on 300 slots, PHASE[ 1 ] is used on 300 slots, PHASE[ 2 ] is used on 300 slots, PHASE[ 3 ] is used on 300 slots, and PHASE[ 4 ] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. 
     Furthermore, for the above-described 750 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16-QAM, PHASE[ 0 ] is used on 150 slots, PHASE[ 1 ] is used on 150 slots, PHASE[ 2 ] is used on 150 slots, PHASE[ 3 ] is used on 150 slots, and PHASE[ 4 ] is used on 150 slots. 
     Further still, for the above-described 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64-QAM, PHASE[ 0 ] is used on 100 slots, PHASE[ 1 ] is used on 100 slots, PHASE[ 2 ] is used on 100 slots, PHASE[ 3 ] is used on 100 slots, and PHASE[ 4 ] is used on 100 slots. 
     As described above, a scheme for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bits making up a single coded block, PHASE[ 0 ] is used on K 0  slots, PHASE[ 1 ] is used on K 1  slots, PHASE[i] is used on K i  slots (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and PHASE[N−1] is used on K N−1  slots, such that Condition #D1-4 is met. 
     (Condition #D1-4) 
     K 0 =K 1  . . . =K i = . . . K N−1 . That is, K a =K b  (for ∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Conditions #D1-4 is preferably met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #D1-4 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #D1-4. 
     (Condition #D1-5) 
     The difference between K a  and K b  satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b) 
       FIG.  35    illustrates the varying numbers of symbols and slots needed in two coded blocks when block codes are used.  FIG.  35    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the transmission device from  FIG.  67    and  FIG.  70   , and the transmission device has two encoders. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  35   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     The transmission device from  FIG.  67    and the transmission device from  FIG.  70    each transmit two streams at once, and have two encoders. As such, the two streams each transmit different code blocks. Accordingly, when the modulation scheme is QPSK, two coded blocks drawn from s 1  and s 2  are transmitted within the same interval, e.g., a first coded block drawn from s 1  is transmitted, then a second coded block drawn from s 2  is transmitted. As such, 3000 slots are needed in order to transmit the first and second coded blocks. 
     By the same reasoning, when the modulation scheme is 16-QAM, 1500 slots are needed to transmit all of the bits making up the two coded blocks, and when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks 
     The following describes the relationship between the above-defined slots and the phase of multiplication, as pertains to methods for a regular change of phase. 
     Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the method for a regular change of phase. That is, the phase changer of the transmission device from  FIG.  67    and  FIG.  70    uses five phase changing values (or phase changing sets) to achieve the period (cycle) of five. (As in  FIG.  69   , five phase changing values are needed in order to perform a change of phase having a period (cycle) of five on switched baseband signal q 2  only. Similarly, in order to perform the change in phase on both switched baseband signals q 1  and q 2 , two phase changing values are needed for each slot. These two phase changing values are termed a phase changing set. Accordingly, here, in order to perform a change of phase having a period (cycle) of five, five such phase changing sets should be prepared). The five phase changing values (or phase changing sets) are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ], PHASE[ 3 ], and PHASE [ 4 ]. 
     For the above-described 3000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is QPSK, PHASE[ 0 ] is used on 600 slots, PHASE[ 1 ] is used on 600 slots, PHASE[ 2 ] is used on 600 slots, PHASE[ 3 ] is used on 600 slots, and PHASE[ 4 ] is used on 600 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 600 times, PHASE[ 1 ] is used on slots 600 times, PHASE[ 2 ] is used on slots 600 times, PHASE[ 3 ] is used on slots 600 times, and PHASE[ 4 ] is used on slots 600 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 600 times, PHASE[ 1 ] is used on slots 600 times, PHASE[ 2 ] is used on slots 600 times, PHASE[ 3 ] is used on slots 600 times, and PHASE[ 4 ] is used on slots 600 times. 
     Similarly, for the above-described 1500 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 16-QAM, PHASE[ 0 ] is used on 300 slots, PHASE[ 1 ] is used on 300 slots, PHASE[ 2 ] is used on 300 slots, PHASE[ 3 ] is used on 300 slots, and PHASE[ 4 ] is used on 300 slots. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 300 times, PHASE[ 1 ] is used on slots 300 times, PHASE[ 2 ] is used on slots 300 times, PHASE[ 3 ] is used on slots 300 times, and PHASE[ 4 ] is used on slots 300 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 300 times, PHASE[ 1 ] is used on slots 300 times, PHASE[ 2 ] is used on slots 300 times, PHASE[ 3 ] is used on slots 300 times, and PHASE[ 4 ] is used on slots 300 times. 
     Similarly, for the above-described 1000 slots needed to transmit the 6000x2 bits making up the two coded blocks when the modulation scheme is 64-QAM, PHASE[ 0 ] is used on 200 slots, PHASE[ 1 ] is used on 200 slots, PHASE[ 2 ] is used on 200 slots, PHASE[ 3 ] is used on 200 slots, and PHASE[ 4 ] is used on 200 slots. 
     Furthermore, in order to transmit the first coded block, PHASE[ 0 ] is used on slots 200 times, PHASE[ 1 ] is used on slots 200 times, PHASE[ 2 ] is used on slots 200 times, PHASE[ 3 ] is used on slots 200 times, and PHASE[ 4 ] is used on slots 200 times. Furthermore, in order to transmit the second coded block, PHASE[ 0 ] is used on slots 200 times, PHASE[ 1 ] is used on slots 200 times, PHASE[ 2 ] is used on slots 200 times, PHASE[ 3 ] is used on slots 200 times, and PHASE[ 4 ] is used on slots 200 times. 
     As described above, a method for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], PHASE[N−2]). As such, in order to transmit all of the bits making up a single coded block, PHASE[ 0 ] is used on K 0  slots, PHASE[ 1 ] is used on K 1  slots, PHASE[i] is used on K i  slots (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and PHASE[N−1] is used on K N−1  slots, such that Condition #D1-6 is met. 
     (Condition #D1-6) 
     K 0 =K 1  . . . =K i = . . . K N−1 . That is, K a =K b  (for ∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b). 
     Further, in order to transmit all of the bits making up the first coded block, PHASE[ 0 ] is used K 0,1  times, PHASE[ 1 ] is used K 1,1  times, PHASE[i] is used K i,1  times (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and PHASE[N−1] is used K N−1,1  times, such that Condition #D1-7 is met.
 
(Condition #D1-7)
 
K 0,1 =K 1,1 = . . . K i,1 = . . . K N−1,1 . That is, K a,1 =K b,1  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b).
 
Furthermore, in order to transmit all of the bits making up the second coded block, PHASE[ 0 ] is used K 0,2  times, PHASE[ 1 ] is used K 1,2  times, PHASE[i] is used K i,2  times (where i=0, 1, 2 . . . N−1 (i being an integer between 0 and N−1)), and PHASE[N−1] is used K N−1,2  times, such that Condition #D1-8 is met.
 
(Condition #D1-8)
 
K 0,2 =K 1,2 = . . . K i,2 = . . . K N−1,2 . That is, K a,2 =K b,2  (∀a and ∀b where a, b,=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b).
 
     Then, when a communication system that supports multiple modulation schemes selects one such supported method for use, Condition #D1-6 Condition #D1-7, and Condition #D1-8 is met for the supported modulation scheme. 
     However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #D1-6 Condition #D1-7, and Condition #D1-8 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #D1-6 Condition #D1-7, and Condition #D1-8. 
     (Condition #D1-9) 
     The difference between Ka and Kb satisfies 0 or 1. That is, |K a −K b | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b) 
     (Condition #D1-10) 
     The difference between K a,1  and K b,1  satisfies 0 or 1. That is, |K a,1 −K b,1 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b) 
     (Condition #D1-11) 
     The difference between K a,2  and K b,2  satisfies 0 or 1. That is, |K a,2 −K b,2 | satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2 . . . N−1 (a and b being integers between 0 and N−1), a≠b) 
     As described above, bias among the phases being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase of multiplication. As such, data reception quality may be improved for the reception device. 
     As described above, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the method for the regular change of phase. As such, N phase changing values (or phase changing sets) PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], and PHASE[N−1] are prepared. However, schemes exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) PHASE[ 0 ], PHASE[ 1 ], PHASE[ 2 ] . . . PHASE[N−2], and PHASE[N−1] may also change the phases of blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement. Although the above examples discuss a phase changing method with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the above-described conditions are satisfied, great quality data reception improvements are realizable for the reception device. 
     Furthermore, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods using a fixed precoding matrix, space-time block coding methods, single-stream transmission, and methods using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission methods. 
     As described in Non-Patent Literature 3, spatial multiplexing MIMO methods involve transmitting signals s 1  and s 2 , which are mapped using a selected modulation scheme, on each of two different antennas. MIMO methods using a fixed precoding matrix involve performing precoding only (with no change in phase). Further, space-time block coding methods are described in Non-Patent Literature 9, 16, and 17. Single-stream transmission methods involve transmitting signal s 1 , mapped with a selected modulation scheme, from an antenna after performing predetermined processing. 
     Schemes using multi-carrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multi-carrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO methods, MIMO methods using a fixed precoding matrix, space-time block coding methods, single-stream transmission, and methods using a regular change of phase may be used. In particular, methods using a regular change of phase on a selected (sub-)carrier group are preferably used to realize the above. 
     Although the present description describes the present Embodiment as a transmission device applying precoding, baseband switching, and change in phase, all of these may be variously combined. In particular, the phase changer discussed for the present Embodiment may be freely combined with the change in phase discussed in all other Embodiments. 
     Embodiment D2 
     The present Embodiment describes a phase change initialization method for the regular change of phase described throughout the present description. This initialization method is applicable to the transmission device from  FIG.  4    when using a multi-carrier method such as OFDM, and to the transmission devices of  FIGS.  67  and  70    when using a single encoder and distributor, similar to  FIG.  4   . 
     The following is also applicable to a method of regularly changing the phase when encoding is performed using block codes as described in Non-Patent Literature 12 through 15, such as QC LDPC Codes (not only QC-LDPC but also LDPC codes may be used), concatenated LDPC and BCH codes, Turbo codes or Duo-Binary Turbo codes using tail-biting, and so on. 
     The following example considers a case where two streams s 1  and s 2  are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information. 
       FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used.  FIG.  34    illustrates the varying numbers of symbols and slots needed in each coded block when block codes are used when, for example, two streams s 1  and s 2  are transmitted as indicated by the above-described transmission device, and the transmission device has only one encoder. (Here, the transmission method may be any single-carrier method or multi-carrier method such as OFDM.) 
     As shown in  FIG.  34   , when block codes are used, there are 6000 bits making up a single coded block. In order to transmit these 6000 bits, the number of required symbols depends on the modulation scheme, being 3000 for QPSK, 1500 for 16-QAM, and 1000 for 64-QAM. 
     Then, given that the above-described transmission device transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s 1  and the other 1500 symbols are assigned to s 2 . As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s 1  and s 2 . 
     By the same reasoning, when the modulation scheme is 16-QAM, 750 slots are needed to transmit all of the bits making up each coded block, and when the modulation scheme is 64-QAM, 500 slots are needed to transmit all of the bits making up each coded block. 
     The following describes a transmission device transmitting modulated signals having a frame configuration illustrated by  FIGS.  71 A and  71 B .  FIG.  71 A  illustrates a frame configuration for modulated signal z 1 ′ or z 1  (transmitted by antenna  312 A) in the time and frequency domains. Similarly,  FIG.  71 B  illustrates a frame configuration for modulated signal z 2  (transmitted by antenna  312 B) in the time and frequency domains. Here, the frequency (band) used by modulated signal z 1 ′ or z 1  and the frequency (band) used for modulated signal z 2  are identical, carrying modulated signals z 1 ′ or z 1  and z 2  at the same time. 
     As shown in  FIG.  71 A , the transmission device transmits a preamble (control symbol) during interval A. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a first and a second coded block. The transmission device transmits the first coded block during interval B. The transmission device then transmits the second coded block during interval C. 
     Further, the transmission device transmits a preamble (control symbol) during interval D. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a third or fourth coded block and so on. The transmission device transmits the third coded block during interval E. The transmission device then transmits the fourth coded block during interval D. 
     Also, as shown in  FIG.  71 B , the transmission device transmits a preamble (control symbol) during interval A. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a first and a second coded block. The transmission device transmits the first coded block during interval B. The transmission device then transmits the second coded block during interval C. 
     Further, the transmission device transmits a preamble (control symbol) during interval D. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a third or fourth coded block and so on. The transmission device transmits the third coded block during interval E. The transmission device then transmits the fourth coded block during interval D. 
       FIG.  72    indicates the number of slots used when transmitting the coded blocks from  FIG.  34   , specifically using 16-QAM as the modulation scheme for the first coded block. Here, 750 slots are needed to transmit the first coded block. 
     Similarly,  FIG.  72    also indicates the number of slots used to transmit the second coded block, using QPSK as the modulation scheme therefor. Here, 1500 slots are needed to transmit the second coded block. 
       FIG.  73    indicates the slots used when transmitting the coded blocks from  FIG.  34   , specifically using QPSK as the modulation scheme for the third coded block. Here, 1500 slots are needed to transmit the coded block. 
     As explained throughout this description, modulated signal z 1 , i.e., the modulated signal transmitted by antenna  312 A, does not undergo a change in phase, while modulated signal z 2 , i.e., the modulated signal transmitted by antenna  312 B, does undergo a change in phase. The following phase changing method is used for  FIGS.  72  and  73   . 
     Before the change in phase can occur, seven different phase changing values must prepared. The seven phase changing values are labelled #0, #1, #2, #3, #4, #5, and #6. The change in phase is regular and periodic. In other words, the phase changing values are applied regularly and periodically, such that the order is #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6 and so on. 
     As shown in  FIG.  72   , given that 750 slots are needed for the first coded block, phase changing value #0 is used initially, such that #0, #1, #2, #3, #4, #5, #6, #0, #1, #2 . . . #3, #4, #5, #6 are used in succession, with the 750th slot using #0 at the final position. 
     The change in phase is then applied to each slot for the second coded block. The present description assumes multi-cast transmission and broadcasting applications. As such, a receiving terminal may have no need for the first coded block and extract only the second coded block. In such circumstances, given that the final slot used for the first coded block uses phase changing value #0, the initial phase changing value used for the second coded block is #1. As such, the following methods are conceivable: 
     (a): The aforementioned terminal monitors the transmission of the first coded block, i.e., monitors the pattern of the phase changing values through the final slot used to transmit the first coded block, and then estimates the phase changing value used for the initial slot of the second coded block; 
     (b): (a) does not occur, and the transmission device transmits information on the phase changing values in use at the initial slot of the second coded block. 
     Scheme (a) leads to greater energy consumption by the terminal due to the need to monitor the transmission of the first coded block. However, scheme (b) leads to reduced data transmission efficiency. 
     Accordingly, there is a need to improve the phase changing value allocation described above. Consider a method in which the phase changing value used to transmit the initial slot of each coded block is fixed. Thus, as indicated in  FIG.  72   , the phase changing value used to transmit the initial slot of the second coded block and the phase changing value used to transmit the initial slot of the first coded block are identical, being #0. 
     Similarly, as indicated in  FIG.  73   , the phase changing value used to transmit the initial slot of the third coded block is not #3, but is instead identical to the phase changing value used to transmit the initial slot of the first and second coded blocks, being #0. 
     As such, the problems accompanying both methods (a) and (b) described above can be constrained while retaining the effects thereof. 
     In the present Embodiment, the method used to initialize the phase changing value for each coded block, i.e., the phase changing value used for the initial slot of each coded block, is fixed so as to be #0. However, other methods may also be used for single-frame units. For example, the phase changing value used for the initial slot of a symbol transmitting information after the preamble or control symbol has been transmitted may be fixed at #0. 
     Embodiment D3 
     The above-described Embodiments discuss a weighting unit using a precoding matrix expressed in complex numbers for precoding. However, the precoding matrix may also be expressed in real numbers. 
     That is, suppose that two baseband signals s 1 ( i ) and s 2 ( i ) (where i is time or frequency) have been mapped (using a modulation scheme), and precoded to obtained precoded baseband signals z 1 ( i ) and z 2 ( i ). As such, mapped baseband signal s 1 ( i ) has an in-phase component of I s1 ( i ) and a quadrature component of Q s1 ( i ), and mapped baseband signal s 2 ( i ) has an in-phase component of I s2 ( i ) and a quadrature component of Q s2 ( i ), while precoded baseband signal z 1 ( i ) has an in-phase component of Iz 1 ( i ) and a quadrature component of Q z1 ( i ), and precoded baseband signal z 2 ( i ) has an in-phase component of I 2 ( i ) and a quadrature component of Q z2 ( i ), which gives the following precoding matrix H r  when all values are real numbers. 
     
       
         
           
             
               
                 
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     Precoding matrix H r  may also be expressed as follows, where all values are real numbers. 
     
       
         
           
             
               
                 
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                             a 
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     where a 11 , a 12 , a 13 , a 14 , a 21 , a 22 , a 23 , a 24 , a 31 , a 32 , a 33 , a 34 , a 41 , a 42 , a 43 , and a 44  are real numbers. However, none of the following may hold: {a 11 =0, a 12 =0, a 13 =0, and a 14 =0}, {a 21 =0, a 22 =0, a 23 =0, and a 24 =0}, {a 31 =0, a 32 =0, a 33 =0, and a 34 =0}, and {a 41 =0, a 42 =0, a 43 =0, and a 44 =0}. Also, none of the following may hold: {a 11 =0, a 21 =0, a 31 =0, and a 41 =0}, {a 12 =0, a 22 =0, a 32 =0, and a 42 =0}, {a 13 =0, a 23 =0, a 33 =0, and a 43 =0}, and {a 14 =0, a 24 =0, a 34 =0, and a 44 =0}. 
     Embodiment E1 
     The present Embodiment describes a transmission scheme as an application of the change in phase to precoded signals (or precoded signals having switched basebands) for a broadcasting system using the DVB-T2 (Digital Video Broadcasting for a second generation digital terrestrial television broadcasting system) standard. First, the configuration of a frame in a broadcasting system using the DVB-T2 standard is described. 
       FIG.  74    illustrates the overall frame configuration of a signal transmitted by a broadcaster using the DVB-T2 standard. Given that DVB-T2 uses an OFDM method, the frame is configured in the time-frequency domain. Thus,  FIG.  74    illustrates frame configuration in the time-frequency domain. The frame includes P1 signalling data ( 7401 ), L1 pre-signalling data ( 7402 ), L1 post-signalling data ( 7403 ), a common PLP (Physical Layer Pipe) ( 7404 ), and PLPs #1 through #N ( 7405 _ 1  through  7405 _N). (Here, L1 pre-signalling data ( 7402 ) and L1 post-signalling data ( 7403 ) are termed P2 symbols.) As such, the P1 signalling data ( 7401 ), L1 pre-signalling data ( 7402 ), L1 post-signalling data ( 7403 ), a common PLP (Physical Layer Pipe) ( 7404 ), and PLPs #1 through #N ( 7405 _ 1  through  7405 _N) form a frame, which is termed a T2 frame, thus constituting a frame configuration unit. 
     The P1 signalling data ( 7401 ) is a symbol used by the reception device for signal detection and frequency synchronization (including frequency offset estimation), that simultaneously serves to transmit information such as the FFT size and whether the modulated signal is transmitted by a SISO or MISO method. (With SISO methods, only one modulated signal is transmitted, while with MISO methods, a plurality of modulated signals are transmitted. In addition, the space-time blocks described in Non-Patent Literature 9, 16, and 17 may be used.) 
     The L1 pre-signalling data ( 7402 ) is used to transmit information regarding the methods used to transmit the frame, concerning the guard interval, the signal processing method information used to reduce the PAPR (Peak-to-Average Power Ratio), the modulation scheme used to transmit the L1 post-signalling data, the FEC method, the coding rate thereof, the length and size of the L1 post-signalling data, them the payload pattern, the cell (frequency region)-specific numbers, and whether normal mode or extended mode is in use (where normal mode and extended mode differ in terms of sub-carrier numbers used to transmit data). 
     The L1 post-signalling data ( 7403 ) is used to transmit such information as the number of PLPs, the frequency region in use, the PLP-specific numbers, the modulation scheme used to transmit the PLPs, the FEC method, the coding rate thereof, the number of blocks transmitted by each PLP, and so on. 
     The common PLP ( 7404 ) and the PLPs #1 through #N ( 7405 _ 1  through  7405 _N) are areas used for data transmission. 
     The frame configuration from  FIG.  74    illustrates the P1 signalling data ( 7401 ), L1 pre-signalling data ( 7402 ), L1 post-signalling data ( 7403 ), the common PLP (Physical Layer Pipe) ( 7404 ), and the PLPs #1 through #N ( 7405 _ 1  through  7405 _N) divided with respect to the time domain for transmission. However, two or more of these signals may occur simultaneously.  FIG.  75    illustrates such a case. As shown, the L1 pre-signalling data, L1 post-signalling data, and common PLP occur at the same timestamp, while PLP # 1  and PLP # 2  occur simultaneously at another timestamp. That is, each signal may coexist at the same point with respect to the time or frequency domain within the frame configuration. 
       FIG.  76    illustrates a sample configuration of a transmission device (e.g., a broadcaster) applying a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals conforming to the DVB-T2 standard. 
     A PLP signal generator  7602  takes PLP transmit data  7601  (data for the PLPs) and a control signal  7609  as input, performs error-correcting coding according to the error-correcting code information for the PLPs included in the control signal  7609  and performs mapping according to the modulation scheme similarly included in the control signal  7609 , and then outputs a PLP (quadrature) baseband signal  7603 . 
     A P2 symbol signal generator  7605  takes P2 symbol transmit data  7604  and the control signal  7609  as input, performs error-correcting coding according to the error-correcting code information for the P2 symbol included in the control signal  7609  and performs mapping according to the modulation scheme similarly included in the control signal  7609 , and then outputs a P2 symbol (quadrature) baseband signal  7606 . 
     A control signal generator  7608  takes P1 symbol transmit data  7607  and the P2 symbol transmit data  7604  as input and outputs the control signal  7609  for the group of symbols from  FIG.  74    (the P1 signalling data ( 7401 ), the L1 pre-signalling data ( 7402 ), the L1 post-signalling data ( 7403 ), the common PLP ( 7404 ), and PLPs #1 through #N ( 7405 _ 1  through  7405 _N)). The control signal  7609  is made up of transmission method information (such as the error-correcting codes and coding rate therefor, the modulation scheme, the block length, the frame configuration, the selected transmission method in which the precoding matrix is regularly changed, the pilot symbol insertion method, IFFT/FFT information, the PAPR reduction method, and the guard interval insertion method) for the symbol group. 
     A frame configurator  7610  takes a PLP baseband signal  7603 , the P2 symbol baseband signal  7606 , and the control signal  7609  as input, performs reordering with respect to the time and frequency domains according to the frame configuration information included in the control signal, and accordingly outputs (quadrature) baseband signal  7611 _ 1  for stream  1  (a mapped signal, i.e., a baseband signal on which the modulation scheme has been used) and (quadrature) baseband signal  7611 _ 2  for stream  2  (also a mapped signal, i.e., a baseband signal on which the modulation scheme has been used). 
     A signal processor  7612  takes the baseband signal for stream  1   7611 _ 1 , the baseband signal for stream  2   7611 _ 2 , and the control signal  7609  as input, and then outputs modulated signals  1  ( 7613 _ 1 ) and  2  ( 7613 _ 2 ), processed according to the transmission method included in the control signal  7609 . 
     Here, the characteristic feature is that when the transmission method for performing the change of phase on precoded (or precoded and switched) signals is selected, the signal processor performs the change in phase on the precoded (or precoded and switched) signals as indicated in  FIGS.  6 ,  25  through  29 , and  69   . The signals so processed are output as processed modulated signal  1  ( 7613 _ 1 ) and processed modulated signal  2  ( 7613 _ 2 ). 
     A pilot inserter  7614 _ 1  takes processed modulated signal  1  ( 7613 _ 1 ) and control signal  7609  as input, inserts pilot symbols into processed modulated signal  1  ( 7613 _ 1 ) according to the pilot symbol insertion method information included in the control signal  7609 , and outputs a post-pilot symbol insertion modulated signal  7615 _ 1 . 
     Another pilot inserter  7614 _ 2  takes processed modulated signal  2  ( 7613 _ 2 ) and control signal  7609  as input, inserts pilot symbols into processed modulated signal  2  ( 7613 _ 2 ) according to the pilot symbol insertion method information included in the control signal  7609 , and outputs a post-pilot symbol insertion modulated signal  7615 _ 2 . 
     An IFFT unit  7616 _ 1  takes post-pilot symbol insertion modulated signal  7615 _ 1  and the control signal  7609  as input, applies an IFFT according to the IFFT method information included in the control signal  7609 , and outputs post-IFFT signal  7617 _ 1 . 
     Another IFFT unit  7616 _ 2  takes post-pilot symbol insertion modulated signal  7615 _ 2  and the control signal  7609  as input, applies an IFFT according to the IFFT method information included in the control signal  7609 , and outputs post-IFFT signal  7617 _ 2 . 
     PAPR reducer  7618 _ 1  takes post-IFFT signal  7617 _ 1  and control signal  7609  as input, applies PAPR-reducing processing to post-IFFT signal  7617 _ 1  according to the PAPR reduction information included in the control signal  7609 , and outputs post-PAPR reduction signal  7619 _ 1 . 
     PAPR reducer  7618 _ 2  takes post-IFFT signal  7617 _ 2  and control signal  7609  as input, applies PAPR-reducing processing to post-IFFT signal  7617 _ 2  according to the PAPR reduction information included in the control signal  7609 , and outputs post-PAPR reduction signal  7619 _ 2 . 
     Guard interval inserter  7620 _ 1  takes post-PAPR reduction signal  7619 _ 1  and the control signal  7609  as input, inserts guard intervals into post-PAPR reduction  7619 _ 1  according to the guard interval insertion method information included in the control signal  7609 , and outputs post-guard interval insertion signal  7621 _ 1 . 
     Guard interval inserter  7620 _ 2  takes post-PAPR reduction signal  7619 _ 2  and the control signal  7609  as input, inserts guard intervals into post-PAPR reduction  7619 _ 2  according to the guard interval insertion method information included in the control signal  7609 , and outputs post-guard interval insertion signal  7621 _ 2 . 
     A P1 symbol inserter  7622  takes the P1 symbol transmit data  7607  and the post-guard interval insertion signals  7621 _ 1  and  7621 _ 2  as input, generates P1 symbol signals from the P1 symbol transmit data  7607 , adds the P1 symbols to the respective post-guard interval insertion signals  7621 _ 1  and  7621 _ 2 , and outputs post-P1 symbol addition signals  7623 _ 1  and  7623 _ 2 . The P1 symbol signals may be added to one or both of post-guard interval insertion signals  7621 _ 1  and  7621 _ 2 . In the former case, the signal to which nothing is added has zero signals as the baseband signal in the interval to which the symbols are added to the other signal. 
     Wireless processor  7624 _ 1  takes post-P1 symbol addition signal  7623 _ 1  as input, performs processing such as frequency conversion and amplification thereon, and outputs transmit signal  7625 _ 1 . Transmit signal  7625 _ 1  is then output as radio waves by antenna  7626 _ 1 . 
     Wireless processor  7624 _ 2  takes post-P1 symbol addition signal  7623 _ 2  as input, performs processing such as frequency conversion and amplification thereon, and outputs transmit signal  7625 _ 2 . Transmit signal  7625 _ 2  is then output as radio waves by antenna  7626 _ 2 . 
       FIG.  77    illustrates a sample frame configuration in the time-frequency domain where a plurality of PLPs are transmitted after the P1 symbol, P2 symbol, and Common PLP have been transmitted. As shown, with respect to the frequency domain, stream  1  (a mapped signal, i.e., a baseband signal on which the modulation scheme has been used) uses sub-carriers #1 through #M, as does stream  2  (also a mapped signal, i.e., a baseband signal on which the modulation scheme has been used). Accordingly, when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at a single frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As shown in  FIG.  77   , interval 1 is used to transmit symbol group  7701  of PLP # 1  using stream s 1  and stream s 2 . Data are transmitted using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix (where no change in phase performed). 
     Interval 2 is used to transmit symbol group  7702  of PLP # 2  using stream s 1 . Data are transmitted using one modulated signal. 
     Interval 3 is used to transmit symbol group  7703  of PLP # 3  using stream s 1  and stream s 2 . Data are transmitted using a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals. 
     Interval 4 is used to transmit symbol group  7704  using stream s 1  and stream s 2 . Data are transmitted using the time-space block codes described in Non-Patent Literature 9, 16, and 17. 
     When a broadcaster transmits PLPs as illustrated by  FIG.  77   , the reception device from  FIG.  77    receiving the transmit signals needs to know the transmission method of each PLP. Accordingly, as described above, the L1 post-signalling data ( 7403  from  FIG.  74   ), being the P2 symbol, should transmit the transmission scheme for each PLP. The following describes an example of a configuration method for P1 and P2 symbols in such circumstances. 
     Table 2 lists specific examples of control information carried by the P1 symbol. 
     
       
         
           
               
               
             
               
                 TABLE 2 
               
               
                   
               
               
                 S1 (3-bit) 
                 Control Information 
               
               
                   
               
             
            
               
                 000 
                 T2_SISO 
               
               
                   
                 (transmission of one modulated signal in the 
               
               
                   
                 DVB-T2 standard) 
               
               
                 001 
                 T2_MISO 
               
               
                   
                 (transmission using time-space block codes  
               
               
                   
                 in the DVB-T2 standard) 
               
               
                 010 
                 NOT_T2 
               
               
                   
                 (using a standard other than DVB-T2) 
               
               
                   
               
            
           
         
       
     
     In the DVB-T2 standard, S1 control information (three bits of data) is used by the reception device to determine whether or not DVB-T2 is being used, and in the affirmative case, to determine the transmission method. 
     As indicated in Table 2, above, the 3-bit S1 data are set to 000 to indicate that the modulated signals being transmitted conform to transmission of one modulated signal in the DVB-T2 standard. 
     Alternatively, the 3-bit S1 data are set to 001 to indicate that the modulated signals being transmitted conform to the use of time-space block codes in the DVB-T2 standard. 
     In DVB-T2, 010 through 111 are reserved for future use. In order to apply the present invention while maintaining compatibility with DVB-T2, the 3-bit S1 data should be set to 010, for example (anything other than 000 and 001 may be used), and should indicate that a standard other than DVB-T2 is being used for the modulated signals. Thus, the reception device or terminal is able to determine that the broadcaster is transmitting using modulated signals conforming to a standard other than DVB-T2 by detecting that the data read 010. 
     The following describes an example of a configuration method for a P2 symbol used when the modulated signals transmitted by the broadcaster conform to a standard other than DVB-T2. In the first example, a scheme of using the P2 symbol within the DVB-T2 standard. 
     Table 3 lists a first example of control information transmitted by the L1 post-signalling data in the P2 symbol. 
     
       
         
           
               
               
             
               
                 TABLE 3 
               
               
                   
               
               
                 PLP_MODE  
                   
               
               
                 (2-bits) 
                 Control Information 
               
               
                   
               
             
            
               
                 00 
                 SISO/SIMO 
               
               
                 01 
                 MISO/MIMO 
               
               
                   
                 (space-time block codes) 
               
               
                 10 
                 MIMO 
               
               
                   
                 (performing a change of phase on precoded signals  
               
               
                   
                 (or precoded signals having switched basebands)) 
               
               
                 11 
                 MIMO 
               
               
                   
                 (using a fixed precoding matrix, or 
               
               
                   
                 using spatial multiplexing) 
               
               
                   
               
            
           
         
       
     
     The above-given tables use the following abbreviations. 
     SISO: Single-Input Single-Output (one modulated signal transmitted and received by one antenna) 
     SIMO: Single-Input Multiple-Output (one modulated signal transmitted and received by multiple antennas) 
     MISO: Multiple-Input Single-Output (multiple modulated signals transmitted by multiple antennas and received by a single antenna) 
     MIMO: Multiple-Input Multiple-Output (multiple modulated signals transmitted and received by multiple antennas) 
     The two-bit data listed in Table 3 are the PLP_MODE information. As shown in  FIG.  77   , this information is control information for informing the terminal of the transmission method (symbol group of PLP # 1  through # 4  in  FIG.  77   ; hereinafter, symbol group). The PLP_MODE information is present in each PLP. That is, in  FIG.  77   , the PLP_MODE information for PLP # 1 , for PLP # 2 , for PLP # 3 , for PLP # 4 , and so on, is transmitted by the broadcaster. Naturally, the terminal acknowledges the transmission method used by the broadcaster for the PLPs by demodulating (or by performing error-correcting decoding on) this information. 
     When the PLP_MODE is set to 00, data are transmitted by that PLP using a method in which a single modulated signal is transmitted. When the PLP_MODE is set to 01, data are transmitted by that PLP using a method in which multiple modulated signals are transmitted using space-time block codes. When the PLP_MODE is set to 10, data are transmitted by that PLP using a method in which a change in phase is performed on precoded (or precoded and switched) signals. When the PLP_MODE is set to 11, data are transmitted by that PLP using a method in which a fixed precoding matrix is used, or in which a spatial multiplexing MIMO system, is used. 
     When the PLP_MODE is set to any of 01 through 11, the broadcaster must transmit the specific processing (e.g., the specific transmission method by which the change in phase is applied to precoded (or precoded and switched) signals, the encoding method of time-space block codes, or the configuration of the precoding matrix) to the terminal. The following describes an alternative to Table 3, as a configuration method for control information that includes the control information necessitated by such circumstances. 
     Table 4 lists a second example of control information transmitted by the L1 post-signalling data in the P2 symbol, different from that of Table 3. 
     
       
         
           
               
               
               
             
               
                 TABLE 4 
               
               
                   
               
               
                   
                 No. of  
                   
               
               
                 Name 
                 bits 
                 Control Information 
               
               
                   
               
             
            
               
                   
               
            
           
           
               
               
               
            
               
                 PLP_MODE  
                 0 
                 SISO/SIMO 
               
               
                 (1-bit) 
                 1 
                 MISO/MIMO, using one of 
               
               
                   
                   
                 (i) space-time block codes; 
               
               
                   
                   
                 (ii) change in phase performed on 
               
               
                   
                   
                 precoded signals (or precoded signals  
               
               
                   
                   
                 having switched basebands); 
               
               
                   
                   
                 (iii) a fixed precoding matrix; and 
               
               
                   
                   
                 (iv) spatial multiplexing 
               
               
                 MIMO_MODE 
                 0 
                 change in phase on precoded signals  
               
               
                 (1-bit) 
                   
                 (or precoded signals having switched  
               
               
                   
                   
                 basebands) is OFF 
               
               
                   
                 1 
                 change in phase on precoded signals  
               
               
                   
                   
                 (or precoded signals having switched  
               
               
                   
                   
                 basebands) is ON 
               
               
                 MIMO_PATTERN#1  
                 00 
                 space-time block codes 
               
               
                 (2-bit) 
                 01 
                 fixed precoding matrix #1 
               
               
                   
                 10 
                 fixed precoding matrix #2 
               
               
                   
                 11 
                 spatial multiplexing 
               
               
                 MIMO_PATTERN#2 
                 00 
                 change in phase on precoded signals  
               
               
                 (2-bit) 
                   
                 (or precoded signals having switched 
               
               
                   
                   
                 basebands), version #1 
               
               
                   
                 01 
                 change in phase on precoded signals  
               
               
                   
                   
                 (or precoded signals having switched  
               
               
                   
                   
                 basebands), version #2 
               
               
                   
                 10 
                 change in phas on precoded signals  
               
               
                   
                   
                 (or precoded signals having switched 
               
               
                   
                   
                 basebands), version #3 
               
               
                   
                 11 
                 change in phase on precoded signals 
               
               
                   
                   
                 (or precoded signals having switched 
               
               
                   
                   
                 basebands), version #4 
               
               
                   
               
            
           
         
       
     
     As indicated in Table 4, four types of control information are possible: 1-bit PLP_MODE information, 1-bit MIMO_MODE information, 2-bit MIMO_PATTERN # 1  information, and 2-bit MIMO_PATTERN # 2  information. As shown in  FIG.  77   , the terminal is notified of the transmission method for each PLP (namely PLP # 1  through # 4 ) by this information. The four types of control information are present in each PLP. That is, in  FIG.  77   , the PLP_MODE information, MIMO_MODE information, MIMO_PATTERN # 1  information, and MIMO_PATTERN # 2  information for PLP # 1 , for PLP # 2 , for PLP # 3 , for PLP # 4 , and so on, is transmitted by the broadcaster. Naturally, the terminal acknowledges the transmission method used by the broadcaster for the PLPs by demodulating (or by performing error-correcting decoding on) this information. 
     When the PLP_MODE is set to 0, data are transmitted by that PLP using a method in which a single modulated signal is transmitted. When the PLP_MODE is set to 1, data are transmitted by that PLP using a method in which any one of the following applies: (i) space-time block codes are used; (ii) a MIMO system is used where a change in phase is performed on precoded (or precoded and switched) signals; (iii) a MIMO system is used where a fixed precoding matrix is used; and (iv) spatial multiplexing is used. 
     When the PLP_MODE is set to 1, the MIMO_MODE information is valid. When the MIMO_MODE information is set to 0, data are transmitted without using a change in phase performed on precoded (or precoded and switched) signals. When the MIMO_MODE information is set to 1, data are transmitted using a change in phase performed on precoded (or precoded signals having switched basebands). 
     When the PLP_MODE is set to 1 and the MIMO_MODE information is set to 0, the MIMO_PATTERN # 1  information is valid. When the MIMO_PATTERN # 1  information is set to 00, data are transmitted using space-time block codes. When the MIMO_PATTERN # 1  information is set to 01, data are transmitted using fixed precoding matrix # 1  for weighting. When the MIMO_PATTERN # 1  information is set to 10, data are transmitted using fixed precoding matrix # 2  for weighting. (Precoding matrix # 1  and precoding matrix # 2  are different matrices.) When the MIMO_PATTERN # 1  information is set to 11, data are transmitted using spatial multiplexing MIMO. 
     When the PLP_MODE is set to 1 and the MIMO_MODE information is set to 1, the MIMO_PATTERN # 2  information is valid. When the MIMO_PATTERN # 2  information is set to 00, data are transmitted using version # 1  of a change in phase on precoded (or precoded signals having switched basebands). When the MIMO_PATTERN # 2  information is set to 01, data are transmitted using version # 2  of a change in phase on precoded (or precoded signals having switched basebands). When the MIMO_PATTERN # 2  information is set to 10, data are transmitted using version # 3  of a change in phase on precoded (or precoded signals having switched basebands). When the MIMO_PATTERN # 2  information is set to 11, data are transmitted using version # 4  of a change in phase on precoded (or precoded signals having switched basebands). Although the change in phase is performed in four different versions #1 through 4, the following three approaches are possible, given two different methods #A and #B: 
     Phase changes performed using method #A and performed using method #B include identical and different changes. 
     A phase changing value included in method #A is not included in method #B; and 
     Multiple phase changes used in method #A are not included in method #B. 
     The control information listed in Table 3 and Table 4, above, is transmitted by the L1 post-signalling data in the P2 symbol. However, in the DVB-T2 standard, the amount of information transmittable as a P2 symbol is limited. Accordingly, the information listed in Tables 3 and 4 is added to the information transmitted by the P2 symbol in the DVB-T2 standard. When this leads to exceeding the limit on information transmittable as the P2 symbol, then as shown in  FIG.  78   , a signalling PLP ( 7801 ) may be prepared in order to transmit necessary control information (at least partially, i.e., transmitting the L1 post-signalling data and the signalling PLP) not included in the DVB-T2 specification. While  FIG.  78    illustrates a frame configuration identical to that of  FIG.  74   , no limitation is intended in this regard. A specific time and specific carrier region may also be allocated in the time-frequency domain for the signalling PLP, as in  FIG.  75   . That is, the signalling PLP may be freely allocated in the time-frequency domain. 
     As described above, selecting a transmission method that uses a multi-carrier method such as OFDM and preserves compatibility with the DVB-T2 standard, and in which the change in phase is performed on precoded (or precoded and switched) signals has the merits of leading to better reception quality in the LOS environment and to greater transmission speeds. While the present invention describes the possible transmission methods for the carriers as being spatial multiplexing MIMO, MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 , no limitation is intended in this manner. 
     Also, although the description indicates that the broadcaster selects one of the aforementioned transmission methods, these are not the only transmission methods available for selection. Other options include: 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 ; 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and space-time block codes; 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmission methods transmitting only stream s 1 ; 
     A transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 ; 
     MIMO using a fixed precoding matrix and a transmission method performing a change of phase on precoded (or on precoded and switched) signals; A transmission method performing a change of phase on precoded (or on precoded and switched) signals and space-time block codes;
 
A transmission method performing a change of phase on precoded (or on precoded and switched) signals and transmission methods transmitting only stream s 1 .
 
As such, by including a transmission method performing a change of phase on precoded (or on precoded and switched) signals, the merits of leading to greater data transmission speeds in the LOS environment and better reception quality for the reception device are achieved.
 
     Here, given that, as described above, S1 needs to be set for the P1 symbol, another configuration method for the control information (regarding the transmission method for each PLP), different from that of Table 3, is possible. For example, Table 5, below. 
     
       
         
           
               
               
             
               
                 TABLE 5 
               
               
                   
               
               
                 PLP_MODE 
                   
               
               
                 (2-bit) 
                 Control Information 
               
               
                   
               
             
            
               
                 00 
                 SISO/SIMO 
               
               
                 01 
                 MISO/MIMO 
               
               
                   
                 (space-time block codes) 
               
               
                 10 
                 MIMO 
               
               
                   
                 (change in phase on precoded signals (or  
               
               
                   
                 precoded signals having switched basebands)) 
               
               
                 11 
                 Reserved 
               
               
                   
               
            
           
         
       
     
     Table 5 differs from Table 3 in that setting the PLP_MODE information to 11 is reserved. As such, when the transmission method for the PLPs is as described in one of the above examples, the number of bits forming the PLP_MODE information as in the examples of Tables 3 and 5 may be made greater or smaller according to the transmission methods available for selection. 
     Similarly, for Table 4, when, for example, a MIMO method is used with a transmission method that does not support changing the phase of precoded signals (or precoded signals having switched basebands), the MIMO_MODE control information is not necessary. Also, when, for example, MIMO schemes using a fixed precoding matrix are not supported, then the MIMO_PATTERN # 1  is not necessary. Also, when multiple precoding matrices are not necessary, 1-bit information may be used instead of 2-bit information. Furthermore, two or more bits may be used when a plurality of precoding matrices are available. 
     The same principles apply to the MIMO_PATTERN # 2  information. When the transmission method does not require a plurality of methods of performing a change of phase on precoded (or precoded and switched) signals, 1-bit information may be used instead of 2-bit information. Furthermore, two or more bits may be used when a plurality of phase changing schemes are available. 
     Furthermore, although the present Embodiment describes a transmission device having two antennas, no limitation is intended in this regard. The control information may also be transmitted using more than two antennas. In such circumstances, the number of bits in each type of control information may be increased as required in order to realize transmission using four antennas. The above description control information transmission in the P1 and P2 symbol also applies to such cases. 
     While  FIG.  77    illustrates the frame configuration for the PLP symbol groups transmitted by the broadcaster as being divided with respect to the time domain, the following variation is also possible. 
     Unlike  FIG.  77   ,  FIG.  79    illustrates an example of a method for arranging the symbols stream s 1  and stream  2  in the time-frequency domain, after the P1 symbol, the P2 symbol, and the Common PLP have been transmitted. In  FIG.  79   , the symbols labelled # 1  are symbols of the symbol group of PLP # 1  from  FIG.  77   . Similarly, the symbols labelled # 2  are symbols of the symbol group of PLP # 2 , the symbols labelled # 3  are symbols of the symbol group of PLP # 3 , and the symbols labelled # 4  are symbols of the symbol group of PLP # 4 , all from  FIG.  77   . As in  FIG.  77   , PLP # 1  is used to transmit data using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix. PLP # 2  is used to transmit data using only one modulated signal. PLP # 3  is used to transmit data using a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals. PLP # 4  is used to transmit data using space-time block codes. 
     In  FIG.  79   , when both s 1  and s 2  have a symbol on the same sub-carrier (given as carrier in  FIG.  79   ) at the same timestamp, a symbol from each of the two streams is present at the common frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As described above,  FIG.  79    differs from  FIG.  77    in that the PLPs are divided with respect to the time domain. In addition,  FIG.  79    has a plurality of PLPs arranged with respect to the time and frequency domains. That is, for example, the symbols of PLP # 1  and PLP # 2  are at timestamp  1 , while the symbols of PLP # 3  and PLP # 4  are at timestamp  3 . As such, PLP symbols having a different index (#X, where X=1, 2, and so on) may be allocated to each symbol (made up of a timestamp and a sub-carrier). 
     Although, for the sake of simplicity,  FIG.  79    lists only #1 and #2 at timestamp  1 , no limitation is intended in this regard. Indices of PLP symbols other than # 1  and # 2  may be at timestamp # 1 . Furthermore, the relationship between PLP indices and sub-carriers at timestamp  1  is not limited to that illustrated by  FIG.  79   . The indices of any PLP symbols may be assigned to any sub-carrier. The same applies to other timestamps, in that the indices of any PLP symbols may be assigned thereto. 
     Unlike  FIG.  77   ,  FIG.  80    illustrates an example of a method for arranging the symbols stream s 1  and stream  2  in the time-frequency domain, after the P1 symbol, the P2 symbol, and the Common PLP have been transmitted. The characteristic feature of  FIG.  80    is that, assuming that using a plurality of antennas for transmission is the basis of the PLP transmission method, then transmission using only stream  1  is not an option for the T2 frame. 
     Accordingly, in  FIG.  80   , PLP symbol group  8001  transmits data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix. Also, symbol group  8002  of PLP # 2  transmits data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals. Further, symbol group  8003  of PLP # 3  transmits data using space-time block code. PLP symbol groups following symbol group  8003  of PLP # 3  transmit data using one of these methods, namely using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix, using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, or using space-time block codes. 
     Unlike  FIG.  79   ,  FIG.  81    illustrates an example of a method for arranging the symbols stream s 1  and stream  2  in the time-frequency domain, after the P1 symbol, the P2 symbol, and the Common PLP have been transmitted. In  FIG.  81   , the symbols labelled # 1  are symbols of the symbol group of PLP # 1  from  FIG.  80   . Similarly, the symbols labelled # 2  are symbols of the symbol group of PLP # 2 , the symbols labelled # 3  are symbols of the symbol group of PLP # 3 , and the symbols labelled # 4  are symbols of the symbol group of PLP # 4 , all from  FIG.  80   . As in  FIG.  80   , PLP # 1  is used to transmit data using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix. PLP # 2  is used to transmit data using a transmission method in which a change of phase is performed on precoded (or precoded and switched) signals. PLP # 3  is used to transmit data using space-time block codes. 
     In  FIG.  81   , when both s 1  and s 2  have a symbol on the same sub-carrier (given as carrier in  FIG.  81   ) at the same timestamp, a symbol from each of the two streams is present at the common frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
       FIG.  81    differs from  FIG.  80    in that the PLPs are divided with respect to the time and frequency domains. That is, for example, the symbols of PLP # 1  and of PLP # 2  are both at timestamp  1 . As such, PLP symbols having a different index (#X, where X=1, 2, and so on) may be allocated to each symbol (made up of a timestamp and a sub-carrier). 
     Although, for the sake of simplicity,  FIG.  81    lists only #1 and #2 at timestamp  1 , no limitation is intended in this regard. Indices of PLP symbols other than # 1  and # 2  may be at timestamp # 1 . Furthermore, the relationship between PLP indices and sub-carriers at timestamp  1  is not limited to that illustrated by  FIG.  81   . The indices of any PLP symbols may be assigned to any sub-carrier. The same applies to other timestamps, in that the indices of any PLP symbols may be assigned thereto. On the other hand, one timestamp may also have symbols of only one PLP assigned thereto, as is the case for timestamp  3 . In other words, any assignment of PLP symbols in the time-frequency domain is allowable. 
     Thus, given that the T2 frame includes no PLPs using transmission methods transmitting only stream s 1 , the dynamic range of the signals received by the terminal may be constrained, which is likely to lead to improved received signal quality. 
     Although  FIG.  81    is described using examples of selecting one of transmitting data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix, transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmitting data using space-time block codes, the selection of transmission method is not limited as such. Other possibilities include: 
     selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, transmitting data using space-time block codes, and transmitting data using a MIMO system using a fixed precoding matrix;
 
selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmitting data using space-time block codes; and
 
selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals and transmitting data using a MIMO system using a fixed precoding matrix.
 
     While the above explanation is given for a T2 frame having multiple PLPs, the following describes a T2 frame having only one PLP. 
       FIG.  82    illustrates a sample frame configuration for stream s 1  and stream s 2  in the time-frequency domain where the T2 frame has only one PLP. Although  FIG.  82    indicates control symbols, these are equivalent to the above-described symbols, such as P1 and P2 symbols. In  FIG.  82   , interval 1 is used to transmit a first T2 frame, interval 2 is used to transmit a second T2 frame, interval 3 is used to transmit a third T2 frame, and interval 4 is used to transmit a fourth T2 frame. 
     Furthermore, the first T2 frame in  FIG.  82    transmits symbol group  8101  of PLP # 1 - 1 . The selected transmission method is spatial multiplexing MIMO or MIMO using a fixed precoding matrix. 
     The second T2 frame transmits symbol group  8102  of PLP # 2 - 1 . The transmission method is transmission using a single modulated signal. 
     The third T2 frame transmits symbol group  8103  of PLP # 3 - 1 . The transmission method is transmission performing a change of phase on precoded (or on precoded and switched) signals. 
     The fourth T2 frame transmits symbol group  8104  of PLP # 4 - 1 . The transmission method is transmission using space-time block codes. 
     In  FIG.  82   , when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at the common frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded signals (or precoded signals having switched basebands), the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As such, the transmission method may be set by taking the data transmission speed and the data reception speed of the terminal into consideration for each PLP. This has the dual merits of allowing the data transmission speed to be enhanced and ensuring high data reception quality. The configuration method for the control information pertaining to the transmission method and so on for the P1 and P2 symbols (and the signalling PLP, where applicable) may be as given by Tables 2 through 5, thus obtaining the same effects.  FIG.  82    differs from  FIG.  77    in that, while the frame configuration from  FIG.  77    and the like includes multiple PLPs in a single T2 frame, thus necessitating control information pertaining to the transmission method and so on of each PLP, the frame configuration of  FIG.  82    includes only one PLP per T2 frame. As such, the only control information needed is for the transmission information and so on pertaining the one PLP. 
     Although the above description discusses methods of transmitting information pertaining to the transmission method of PLPs using P1 and P2 symbols (and the signalling PLP, where applicable), the following describes a method of transmitting information pertaining to the transmission method of PLPs without using the P2 symbol. 
       FIG.  83    illustrates a frame configuration in the time-frequency domain applicable when a terminal receiving data transmitted by a broadcaster is not compatible with the DVB-T2 standard. In  FIG.  83   , components operating in the manner described for  FIG.  74    use identical reference numbers. The frame of  FIG.  83    includes P1 signalling data ( 7401 ), first signalling data ( 8301 ), second signalling data ( 8302 ), a common PLP ( 7404 ), and PLPs # 1  through #N ( 7405 _ 1  through  7405 _N). As such, the P1 signalling data ( 7401 ), the first signalling data ( 8301 ), the second signalling data ( 8302 ), the common PLP ( 7404 ), and the PLPs # 1  through #N ( 7405 _ 1  through  7405 _N) form a frame, thus constituting a frame unit. 
     The P1 signalling data ( 7401 ) are a symbol used for signal reception by the reception device and for frequency synchronization (including frequency offset estimation). In addition, these data transmit identification regarding whether or not the frame conforms to the DVB-T2 standard, e.g., using the S1 data as indicated in Table 2 for this purpose. 
     The first signalling data ( 8301 ) are used to transmit information regarding the methods used to transmit the frame, concerning the guard interval, the signal processing method information used to reduce the PAPR, the modulation scheme used to transmit the L1 post-signalling data, the FEC method, the coding rate thereof, the length and size of the L1 post-signalling data, them the payload pattern, the cell (frequency region)-specific numbers, and whether normal mode or extended mode is in use, and other such information. Here, the first signalling data ( 8301 ) need not necessarily be data conforming to the DVB-T2 standard. 
     The second signalling data ( 8302 ) is used to transmit such information as the number of PLPs, the frequency region in use, the PLP-specific numbers, the modulation scheme used to transmit the PLPs, the FEC method, the coding rate thereof, the number of blocks transmitted by each PLP, and so on. 
     The frame configuration from  FIG.  83    illustrates the first signalling data ( 8301 ), the second signalling data ( 8302 ), the L1 post-signalling data ( 7403 ), the common PLP ( 7404 ), and the PLPs # 1  through #N ( 7405 _ 1  through  7405 _N) divided with respect to the time domain for transmission. However, two or more of these signals may occur simultaneously.  FIG.  84    illustrates such a case. As shown in  FIG.  84   , the first signalling data, the second signalling data, and the common PLP share a common timestamp, while PLP # 1  and PLP # 2  share a different common timestamp. That is, each signal may coexist at the same point with respect to the time or frequency domain within the frame configuration. 
       FIG.  85    illustrates a sample configuration of a transmission device (e.g., a broadcaster) applying a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals as explained thus far, but conforming to a standard other than the DVB-T2 standard. In  FIG.  85   , components operating in the manner described for  FIG.  76    use identical reference numbers and invoke the above descriptions. 
     A control signal generator  7608  takes first and second signalling data  8501  and P1 symbol transmit data  7607  as input, and outputs the control signal  7609  (made up of such information as the error-correcting codes and coding rate therefor, the modulation scheme, the block length, the frame configuration, the selected transmission method in which the precoding matrix is regularly changed, the pilot symbol insertion method, IFFT/FFT information, the PAPR reduction method, and the guard interval insertion method) for the transmission method of each symbol group of  FIG.  83   . 
     A control symbol signal generator  8502  takes the first and second signalling data transmit data  8501  and the control signal  7609  as input, performs error-correcting coding according to the error-correcting code information for the first and second signalling data included in the control signal  7609  and performs mapping according to the modulation scheme similarly included in the control signal  7609 , and then outputs a first and second signalling data (quadrature) baseband signal  8503 . 
     In  FIG.  85   , the frame configurator  7610  takes the baseband signal  8503  generated by the control symbol signal generator  8502  as input, rather than the baseband signal  7606  generated by the P2 symbol signal generator  7605  from  FIG.  76   . 
     The following describes, with reference to  FIG.  77   , a transmission method for control information (information transmitted by the P1 symbol and by the first and second signalling data) and for the frame configuration of the transmit signal for a broadcaster (base station) applying a transmission method in which a change in phase is performed on precoded (or on precoded and switched) signals in a system not conforming to the DVB-T2 standard. 
       FIG.  77    illustrates a sample frame configuration in the time-frequency domain where a plurality of PLPs are transmitted after the first and second signalling data and the Common PLP have been transmitted. In  FIG.  77   , stream s 1  uses sub-carrier # 1  through sub-carrier #M in the frequency domain. Similarly, stream s 2  also uses sub-carrier # 1  through sub-carrier #M in the frequency domain. Accordingly, when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at a single frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As shown in  FIG.  77   , interval 1 is used to transmit symbol group  7701  of PLP # 1  using stream s 1  and stream s 2 . Data are transmitted using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix. 
     Interval 2 is used to transmit symbol group  7702  of PLP # 2  using stream s 1 . Data are transmitted using one modulated signal. 
     Interval 3 is used to transmit symbol group  7703  of PLP # 3  using stream s 1  and stream s 2 . Data are transmitted using a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals. 
     Interval 4 is used to transmit symbol group  7704  of PLP # 4  using stream s 1  and stream s 2 . Data are transmitted using the time-space block codes. 
     When a broadcaster transmits PLPs as illustrated by  FIG.  77   , the reception device from  FIG.  64    receiving the transmit signals needs to know the transmission method of each PLP. Accordingly, as described above, the first and second signalling data are used to transmit the transmission method for each PLP. The following describes an example of a configuration method for the P1 symbol and for the first and second signalling data in such circumstances. A specific example of control information carried by the P1 symbol is given in Table 2. 
     In the DVB-T2 standard, S1 control information (three bits of data) is used by the reception device to determine whether or not DVB-T2 is being used, and in the affirmative case, to determine the transmission method. The 3-bit S1 data are set to 000 to indicate that the modulated signals being transmitted conform to transmission of one modulated signal in the DVB-T2 standard. 
     Alternatively, the 3-bit S1 data are set to 001 to indicate that the modulated signals being transmitted conform to the use of time-space block codes in the DVB-T2 standard. 
     In DVB-T2, 010 through 111 are reserved for future use. In order to apply the present invention while maintaining compatibility with DVB-T2, the 3-bit S1 data should be set to 010, for example (anything other than 000 and 001 may be used), and should indicate that a standard other than DVB-T2 is being used for the modulated signals. Thus, the reception device or terminal is able to determine that the broadcaster is transmitting using modulated signals conforming to a standard other than DVB-T2 by detecting that the data read 010. 
     The following describes a configuration method for the first and second signalling data used when the modulated signals transmitted by the broadcaster do not conform to the DVB-T2 standard. A second example of control information for the first and second signalling data is given by Table 3. 
     The two-bit data listed in Table 3 are the PLP_MODE information. As shown in  FIG.  77   , this information is control information for informing the terminal of the transmission method for each PLP (PLP # 1  through # 4  in  FIG.  77   ). The PLP_MODE information is present in each PLP. That is, in  FIG.  77   , the PLP_MODE information for PLP # 1 , for PLP # 2 , for PLP # 3 , for PLP # 4 , and so on, is transmitted by the broadcaster. Naturally, the terminal acknowledges the transmission method used by the broadcaster for the PLPs by demodulating (or by performing error-correcting decoding on) this information. 
     When the PLP_MODE is set to 00, data are transmitted by that PLP using a method in which a single modulated signal is transmitted. When the PLP_MODE is set to 01, data are transmitted by that PLP using a method in which multiple modulated signals are transmitted using space-time block codes. When the PLP_MODE is set to 10, data are transmitted by that PLP using a method in which a change in phase is performed on precoded (or precoded and switched) signals. When the PLP_MODE is set to 11, data are transmitted by that PLP using a method in which a fixed precoding matrix is used, or in which a spatial multiplexing MIMO system, is used. 
     When the PLP_MODE is set to any of 01 through 11, the broadcaster must transmit the specific processing (e.g., the specific transmission method by which a change in phase is applied to precoded (or precoded and switched) signals, the encoding method of time-space block codes, or the configuration of the precoding matrix) to the terminal. The following describes an alternative to Table 3, as a configuration method for control information that includes the control information necessitated by such circumstances. 
     A second example of control information for the first and second signalling data is given by Table 4. 
     As indicated in Table 4, four types of control information are possible: 1-bit PLP_MODE information, 1-bit MIMO_MODE information, 2-bit MIMO_PATTERN # 1  information, and 2-bit MIMO_PATTERN # 2  information. As shown in  FIG.  77   , the terminal is notified of the transmission method for each PLP (namely PLP # 1  through # 4 ) by this information. The four types of control information are present in each PLP. That is, in  FIG.  77   , the PLP_MODE information, MIMO_MODE information, MIMO_PATTERN # 1  information, and MIMO_PATTERN # 2  information for PLP # 1 , for PLP # 2 , for PLP # 3 , for PLP # 4 , and so on, is transmitted by the broadcaster. Naturally, the terminal acknowledges the transmission method used by the broadcaster for the PLPs by demodulating (or by performing error-correcting decoding on) this information. 
     When the PLP_MODE is set to 0, data are transmitted by that PLP using a method in which a single modulated signal is transmitted. When the PLP_MODE is set to 1, data are transmitted by that PLP using a method in which any one of the following applies: (i) space-time block codes are used; (ii) a MIMO system is used where a change in phase is performed on precoded (or precoded and switched) signals; (iii) a MIMO system is used where a fixed precoding matrix is used; and (iv) spatial multiplexing is used. 
     When the PLP_MODE is set to 1, the MIMO_MODE information is valid. When the MIMO_MODE information is set to 0, data are transmitted without using a change in phase performed on recoded signals (or precoded signals having switched basebands). When the MIMO_MODE information is set to 1, data are transmitted using a change in phase performed on recoded signals (or precoded signals having switched basebands). 
     When the PLP_MODE information is set to 1 and the MIMO_MODE information is set to 0, the MIMO_PATTERN # 1  information is valid. As such, when the MIMO_PATTERN # 1  information is set to 00, data are transmitted using space-time block codes. When the MIMO_PATTERN # 1  information is set to 01, data are transmitted using fixed precoding matrix # 1  for weighting. When the MIMO_PATTERN # 1  information is set to 10, data are transmitted using fixed precoding matrix # 2  for weighting. (Precoding matrix # 1  and precoding matrix # 2  are different matrices.) When the MIMO_PATTERN # 1  information is set to 11, data are transmitted using spatial multiplexing MIMO. 
     When the PLP_MODE information is set to 1 and the MIMO_MODE information is set to 1, the MIMO_PATTERN # 2  information is valid. When the MIMO_PATTERN # 2  information is set to 00, data are transmitted using version # 1  of a change in phase on precoded (or precoded and switched) signals. When the MIMO_PATTERN # 2  information is set to 01, data are transmitted using version # 2  of a change in phase on precoded (or precoded signals having switched basebands). When the MIMO_PATTERN # 2  information is set to 10, data are transmitted using version # 3  of a change in phase on precoded (or precoded signals having switched basebands). When the MIMO_PATTERN # 2  information is set to 11, data are transmitted using version # 4  of a change in phase on precoded (or precoded signals having switched basebands). Although the change in phase is performed in four different versions #1 through 4, the following three approaches are possible, given two different methods #A and #B: 
     Phase changes performed using method #A and performed using method #B include identical and different changes. 
     Some phase changing values are included in method #A but are not included in method #B; and 
     Multiple phase changes used in method #A are not included in method #B. 
     The control information listed in Table 3 and Table 4, above, is transmitted by the first and second signalling data. In such circumstances, there is no particular need to use the PLPs to transmit the control information. 
     As described above, selecting a transmission method that uses a multi-carrier method such as OFDM while being identifiable as differing from the DVB-T2 standard, and in which a change of phase is performed on precoded (or precoded and switched) signals has the merits of leading to better reception quality in the LOS environment and to greater transmission speeds. While the present invention describes the possible transmission methods for the carriers as being spatial multiplexing MIMO, MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 , no limitation is intended in this manner. 
     Also, although the description indicates that the broadcaster selects one of the aforementioned transmission methods, these are not the only transmission methods available for selection. Other options include: 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 ; 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and space-time block codes; 
     MIMO using a fixed precoding matrix, a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmission methods transmitting only stream s 1 ; 
     A transmission method performing a change of phase on precoded (or on precoded and switched) signals, space-time block codes, and transmission methods transmitting only stream s 1 ; 
     MIMO using a fixed precoding matrix and a transmission method performing a change of phase on precoded (or on precoded and switched) signals; 
     A transmission method performing a change of phase on precoded (or on precoded and switched) signals and space-time block codes; and 
     A transmission method performing a change of phase on precoded (or on precoded and switched) signals and transmission methods transmitting only stream s 1 . 
     As such, by including a transmission method performing a change of phase on precoded (or on precoded and switched) signals, the merits of leading to greater data transmission speeds in the LOS environment and better reception quality for the reception device are achieved. 
     Here, given that, as described above, the S1 data is set for the P1 symbol, another configuration method for the control information (regarding the transmission method for each PLP) transmitted as the first and second signalling data, different from that of Table 3, is possible. For example, see Table 5, above. 
     Table 5 differs from Table 3 in that setting the PLP_MODE information to 11 is reserved. As such, when the transmission method for the PLPs is as described in one of the above examples, the number of bits forming the PLP_MODE information as in the examples of Tables 3 and 5 may be made greater or smaller according to the transmission methods available for selection. 
     Similarly, for Table 4, when, for example, a MIMO method is used with a transmission method that does not support changing the phase of precoded (or precoded and switched) signals, the MIMO_MODE control information is not necessary. Also, when, for example, MIMO schemes using a fixed precoding matrix are not supported, then the MIMO_PATTERN # 1  is not necessary. Also, when multiple precoding matrices are not necessary, 1-bit information may be used instead of 2-bit information. Furthermore, two or more bits may be used when a plurality of precoding matrices are available. 
     The same principles apply to the MIMO_PATTERN # 2  information. When the transmission schemes does not require a plurality of methods of performing a change of phase on precoded (or precoded and switched) signals, 1-bit information may be used instead of 2-bit information. Furthermore, two or more bits may be used when a plurality of phase changing schemes are available. 
     Furthermore, although the present Embodiment describes a transmission device having two antennas, no limitation is intended in this regard. The control information may also be transmitted using more than two antennas. In such circumstances, the number of bits in each type of control information may be increased as required in order to realize transmission using four antennas. The above description control information transmission in the P1 symbol and in the first and second signalling data also applies to such cases. 
     While  FIG.  77    illustrates the frame configuration for the PLP symbol groups transmitted by the broadcaster as being divided with respect to the time domain, the following variation is also possible. 
     Unlike  FIG.  77   ,  FIG.  79    illustrates an example of a method for arranging the symbols stream s 1  and stream  2  in the time-frequency domain, after the P1 symbol, the first and second signalling data, and the Common PLP have been transmitted. 
     In  FIG.  79   , the symbols labelled # 1  are symbols of the symbol group of PLP # 1  from  FIG.  77   . Similarly, the symbols labelled # 2  are symbols of the symbol group of PLP # 2 , the symbols labelled # 3  are symbols of the symbol group of PLP # 3 , and the symbols labelled # 4  are symbols of the symbol group of PLP # 4 , all from  FIG.  77   . As in  FIG.  77   , PLP # 1  is used to transmit data using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix. PLP # 2  is used to transmit data using only one modulated signal. PLP # 3  is used to transmit data using a transmission method in which a change in phase is performed on precoded (or precoded and switched) signals. PLP # 4  is used to transmit data using space-time block codes. 
     In  FIG.  79   , when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at the common frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As described above,  FIG.  79    differs from  FIG.  77    in that the PLPs are divided with respect to the time domain. In addition,  FIG.  79    has a plurality of PLPs arranged with respect to the time and frequency domains. That is, for example, the symbols of PLP # 1  and PLP # 2  are at timestamp  1 , while the symbols of PLP # 3  and PLP # 4  are at timestamp  3 . As such, PLP symbols having a different index (#X, where X=1, 2, and so on) may be allocated to each symbol (made up of a timestamp and a sub-carrier). 
     Although, for the sake of simplicity,  FIG.  79    lists only #1 and #2 at timestamp  1 , no limitation is intended in this regard. Indices of PLP symbols other than # 1  and # 2  may be at timestamp # 1 . Furthermore, the relationship between PLP indices and sub-carriers at timestamp  1  is not limited to that illustrated by  FIG.  79   . The indices of any PLP symbols may be assigned to any sub-carrier. The same applies to other timestamps, in that the indices of any PLP symbols may be assigned thereto. 
     Unlike  FIG.  77   ,  FIG.  80    illustrates an example of a method for arranging the symbols stream s 1  and stream s 2  in the time-frequency domain, after the P1 symbol, the first and second signalling data, and the Common PLP have been transmitted. The characteristic feature of  FIG.  80    is that, assuming that using a plurality of antennas for transmission is the basis of the PLP transmission method, then transmission using only stream  1  is not an option for the T2 frame. 
     Accordingly, in  FIG.  80   , PLP symbol group  8001  transmits data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix. Also, symbol group  8002  of PLP # 2  transmits data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals. Further, symbol group  8003  of PLP # 3  transmits data using space-time block code. PLP symbol groups following symbol group  8003  of PLP # 3  transmit data using one of these methods, namely using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix, using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, or using space-time block codes. 
     Unlike  FIG.  79   ,  FIG.  81    illustrates an example of a method for arranging the symbols stream s 1  and stream s 2  in the time-frequency domain, after the P1 symbol, the first and second signalling data, and the Common PLP have been transmitted. 
     In  FIG.  81   , the symbols labelled # 1  are symbols of the symbol group of PLP # 1  from  FIG.  80   . Similarly, the symbols labelled # 2  are symbols of the symbol group of PLP # 2 , the symbols labelled # 3  are symbols of the symbol group of PLP # 3 , and the symbols labelled # 4  are symbols of the symbol group of PLP # 4 , all from  FIG.  80   . As in  FIG.  80   , PLP # 1  is used to transmit data using a spatial multiplexing MIMO system as illustrated by  FIG.  23   , or by using a MIMO system with a fixed precoding matrix. PLP # 2  is used to transmit data using a transmission method in which a change of phase is performed on precoded (or precoded and switched) signals. PLP # 3  is used to transmit data using space-time block codes. 
     In  FIG.  81   , when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at the common frequency. As explained in other Embodiments, when using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As described above,  FIG.  81    differs from  FIG.  80    in that the PLPs are divided with respect to the time domain. In addition,  FIG.  81    has a plurality of PLPs arranged with respect to the time and frequency domains. That is, for example, the symbols of PLP # 1  and of PLP # 2  are both at timestamp  1 . As such, PLP symbols having a different index (#X, where X=1, 2, and so on) may be allocated to each symbol (made up of a timestamp and a sub-carrier). 
     Although, for the sake of simplicity,  FIG.  81    lists only #1 and #2 at timestamp  1 , no limitation is intended in this regard. Indices of PLP symbols other than # 1  and # 2  may be at timestamp # 1 . Furthermore, the relationship between PLP indices and sub-carriers at timestamp  1  is not limited to that illustrated by  FIG.  81   . The indices of any PLP symbols may be assigned to any sub-carrier. The same applies to other timestamps, in that the indices of any PLP symbols may be assigned thereto. On the other hand, one timestamp may also have symbols of only one PLP assigned thereto, as is the case for timestamp  3 . In other words, any assignment of PLP symbols in the time-frequency domain is allowable. 
     Thus, given that the frame unit includes no PLPs using transmission methods transmitting only stream s 1 , the dynamic range of the signals received by the terminal may be constrained, which is likely to lead to improved received signal quality 
     Although  FIG.  81    is described using examples of selecting one of transmitting data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix, transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmitting data using space-time block codes, the selection of transmission method is not limited as such. Other possibilities include: 
     selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, transmitting data using space-time block codes, and transmitting data using a MIMO system using a fixed precoding matrix;
 
selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals, and transmitting data using space-time block codes; and
 
selecting one of transmitting data using a transmission method performing a change of phase on precoded (or on precoded and switched) signals and transmitting data using a MIMO system using a fixed precoding matrix.
 
     While the above explanation is given for a frame unit having multiple PLPs, the following describes a frame unit having only one PLP. 
       FIG.  82    illustrates a sample frame configuration for stream s 1  and stream s 2  in the time-frequency domain where the frame unit has only one PLP. 
     Although  FIG.  82    indicates control symbols, these are equivalent to the above-described P1 symbol and to the first and second signalling data. In  FIG.  82   , interval 1 is used to transmit a first frame unit, interval 2 is used to transmit a second frame unit, interval 3 is used to transmit a third frame unit, and interval 4 is used to transmit a fourth frame unit. 
     Furthermore, the first frame unit in  FIG.  82    transmits symbol group  8101  of PLP # 1 - 1 . The transmission method is spatial multiplexing MIMO or MIMO using a fixed precoding matrix. 
     The second frame unit transmits symbol group  8102  of PLP # 2 - 1 . The transmission method is transmission using a single modulated signal. 
     The third frame unit transmits symbol group  8103  of PLP # 3 - 1 . The transmission method is a transmission method performing a change of phase on precoded (or on precoded and switched) signals. 
     The fourth frame unit transmits symbol group  8104  of PLP # 4 - 1 . The transmission method is transmission using space-time block codes. 
     In  FIG.  82   , when both s 1  and s 2  have a symbol on the same sub-carrier at the same timestamp, a symbol from each of the two streams is present at the common frequency. When using a transmission method that involves performing a change of phase on precoded (or precoded and switched) signals, the change in phase may be performed in addition to weighting using the precoding matrix (and, where applicable, after switching the baseband signal). Accordingly, signals z 1  and z 2  are obtained. The signals z 1  and z 2  are each output by a different antenna. 
     As such, the transmission method may be set by taking the data transmission speed and the data reception speed of the terminal into consideration for each PLP. This has the dual merits of allowing the data transmission speed to be enhanced and ensuring high data reception quality. The configuration method for the control information pertaining to the transmission method and so on for the P1 symbol and for the first and second signalling data may be as given by Tables 2 through 5, thus obtaining the same effects. The frame configuration of  FIG.  82    differs from that of  FIG.  77    and the like, where each frame unit has multiple PLPs, and control information pertaining to the transmission method for each of the PLPs is required. In  FIG.  82   , each frame unit has only one PLP, and thus, the only control information needed is for the transmission information and so on pertaining to that single PLP. 
     The present Embodiment describes a method applicable to a system using a DVB standard and in which the transmission method involves performing a change of phase on precoded (or precoded and switched) signals. The transmission method involving performing a change of phase on precoded signals (or precoded signals having switched basebands) is described in the present description. Although the present Embodiment uses “control symbol” as a term of art, this term has no influence on the present invention. 
     The following describes the space-time block codes discussed in the present description and included in the present Embodiment. 
       FIG.  94    illustrates the configuration of a modulated signal using space-time block codes. As shown, a space-time block coder ( 9402 ) takes a baseband signal based on a modulated signal as input. For example, the space-time block coder ( 9402 ) takes symbol s 1 , symbol s 2 , and so on as input. Then, as shown in  FIG.  94   , space-time block coding is performed, resulting in z 1  ( 9403 A) taking s 1  as symbol # 0 , −s 2 * as symbol # 1 , s 3  as symbol # 2 , −s 4 * as symbol # 3 , and so on, and z 2  ( 9403 B) taking s 2  as symbol # 0 , s 1 * as symbol # 1 , s 4  as symbol # 2 , s 3 * as symbol # 3 , and so on. Here, symbol #X of z 1  and symbol #X of z 2  are simultaneous signals on a common frequency, each broadcast from a different antenna. The arrangement of symbols in the space-time block codes is not restricted to the time domain. A group of symbols may also be arranged in the frequency domain, or in the time-frequency domain, as required. Furthermore, the space-time block coding method of  FIG.  94    is given as an example of space-time block codes. Other space-time block codes may also be applied to each Embodiment discussed in the present description. 
     Embodiment E2 
     The present Embodiment describes a reception method and a reception device applicable to a communication system using the DVB-T2 standard when the transmission method described in Embodiment E1, which involves performing a change of phase on precoded (or on precoded and switched) signals, is used. 
       FIG.  86    illustrates a sample configuration for a reception device in a terminal, for use when the transmission device of the broadcaster from  FIG.  76    applies a transmission method involving a change in phase of precoded (or precoded and switched) signals. Components thereof operating identically to those of  FIG.  7    use the same reference numbers thereas. 
     In  FIG.  86   , a P1 symbol detector and decoder  8601  receives the signal transmitted by the broadcaster and takes baseband signals  704 _X and  704 _Y as input, thereby performing signal detection and frequency synchronization. The P1 symbol detector and decoder  8601  simultaneously obtains the control information included in the P1 symbol (by performing demodulation and error-correcting decoding thereon) and outputs the P1 symbol control information  8602  so obtained. 
     OFDM-related processors  8600 _X and  8600 _Y take the P1 symbol control information  8602  as input and modify the OFDM signal processing method (such as the Fourier transform) accordingly. (This is possible because, as described in Embodiment E1, the signals transmitted by the broadcaster include transmission method information in the P1 symbol.) The OFDM-related processors  8600 _X and  8600 _Y then output the baseband signals  704 _X and  704 _Y after performing demodulation thereon according to the signal processing method. 
     A P2 symbol demodulator  8603  (which may also apply to the signalling PLP) takes the baseband signals  704 _X and  704 _Y and the P1 symbol control information  8602  as input, performs signal processing and demodulation (including error-correcting decoding) in accordance with the P1 symbol control information, and outputs P2 symbol control information  8604 . 
     A control information generator  8605  takes the P1 symbol control information  8602  and the P2 symbol control information  8604  as input, bundles the control information (pertaining to reception operations), and outputs a control signal  8606 . Then, as shown in  FIG.  86   , the control signal  8606  is input to each component. 
     A signal processor  711  takes signals  706 _ 1 ,  706 _ 2 ,  708 _ 1 ,  708 _ 2 ,  704 _X, and  704 _Y, as well as control signal  8606 , as input, performs demodulation an decoding according to the information included in the control signal  8606 , and outputs received data  712 . The information included in the control signal pertains to the transmission method, modulation scheme, error-correcting coding method and coding rate thereof, error-correcting code block size, and so on used for each PLP. 
     When the transmission method used for the PLPs is one of spatial multiplexing MIMO, MIMO using a fixed precoding matrix, and a transmission method performing a change of phase on precoded (or on precoded and switched) signals, demodulation is performed by obtaining received (baseband) signals using the output of the channel estimators ( 705 _ 1 ,  705 _ 2 ,  707 _ 1 , and  707 _ 2 ) and the relationship of the received (baseband) signals to the transmit signals. When the transmission method involves performing a change of phase on precoded (or precoded and switched) signals, demodulation is performed using the output of the channel estimators ( 705 _ 1 ,  705 _ 2 ,  707 _ 1 , and  707 _ 2 ), the received (baseband) signals, and the relationship given by Math. 48 (formula 48). 
       FIG.  87    illustrates a sample configuration for a reception device in a terminal, for use when the transmission device of the broadcaster from  FIG.  85    applies a transmission method involving a change in phase of precoded (or precoded and switched) signals. Components thereof operating identically to those of  FIGS.  7  and  86    use the same reference numbers thereas. 
     The reception device from  FIG.  87    differs from that of  FIG.  86    in that, while the latter receives data from signals conforming to the DVB-T2 standard and to other standards, the former receives data only from signals conforming to a standard other than DVB-T2. 
     In  FIG.  87   , a P1 symbol detector and decoder  8601  receives the signal transmitted by the broadcaster and takes baseband signals  704 _X and  704 _Y as input, thereby performing signal detection and frequency synchronization. The P1 symbol detector and decoder  8601  simultaneously obtains the control information included in the P1 symbol (by performing demodulation and error-correcting decoding thereon) and outputs the P1 symbol control information  8602  so obtained. 
     OFDM-related processors  8600 _X and  8600 _Y take the P1 symbol control information  8602  as input and modify the OFDM signal processing method accordingly. (This is possible because, as described in Embodiment E1, the signals transmitted by the broadcaster include transmission method information in the P1 symbol.) The OFDM-related processors  8600 _X and  8600 _Y then output the baseband signals  704 _X and  704 _Y after performing demodulation thereon according to the signal processing method. 
     A first and second signalling data demodulator  8701  (which may also apply to the signalling PLP) takes the baseband signals  704 _X and  704 _Y and the P1 symbol control information  8602  as input, performs signal processing and demodulation (including error-correcting decoding) in accordance with the P1 symbol control information, and outputs first and second signalling data control information  8702 . 
     A control information generator  8605  takes the P1 symbol control information  8602  and the first and second signalling data control information  8702  as input, bundles the control information (pertaining to reception operations), and outputs a control signal  8606 . Then, as shown in  FIG.  86   , the control signal  8606  is input to each component. 
     A signal processor  711  takes signals  706 _ 1 ,  706 _ 2 ,  708 _ 1 ,  708 _ 2 ,  704 _X, and  704 _Y, as well as control signal  8606 , as input, performs demodulation an decoding according to the information included in the control signal  8606 , and outputs received data  712 . The information included in the control signal pertains to the transmission method, modulation scheme, error-correcting coding method and coding rate thereof, error-correcting code block size, and so on used for each PLP. 
     When the transmission method used for the PLPs is one of spatial multiplexing MIMO, MIMO using a fixed precoding matrix, and a transmission method performing a change of phase on precoded (or on precoded and switched) signals, demodulation is performed by obtaining received (baseband) signals using the output of the channel estimators ( 705 _ 1 ,  705 _ 2 ,  707 _ 1 , and  707 _ 2 ) and the relationship of the received (baseband) signals to the transmit signals. When the transmission method involves performing a change of phase on precoded (or precoded and switched) signals, demodulation is performed using the output of the channel estimators ( 705 _ 1 ,  705 _ 2 ,  707 _ 1 , and  707 _ 2 ), the received (baseband) signals, and the relationship given by Math. 48 (formula 48). 
       FIG.  88    illustrates the configuration of a reception device for a terminal compatible with the DVB-T2 standard and with standards other than DVB-T2. Components thereof operating identically to those of  FIGS.  7  and  86    use the same reference numbers thereas. 
       FIG.  88    differs from  FIGS.  86  and  87    in that the reception device of the former is compatible with signals conforming to the DVB-T2 standard as well as signals conforming to other standards. As such, the reception device includes a P2 symbol or first and second signalling data demodulator  8801 , in order to enable demodulation. 
     The P2 symbol or first and second signalling data demodulator  8801  takes the baseband signals  704 _X and  704 _Y, as well as the P1 symbol control information  8602 , as input, uses the P1 symbol control information to determine whether the received signals conform to the DVB-T2 standard or to another standard (e.g., using Table in such a determination), performs signal processing and demodulation (including error-correcting decoding), and outputs control information  8802 , which includes information indicating the standard to which the received signals conform. Otherwise, the operations are identical to those explained for  FIGS.  86  and  87   . 
     A reception device configured as described in the above Embodiment and receiving signals transmitted by a broadcaster having the transmission device described in Embodiment E1 provides higher received data quality by applying appropriate signal processing. In particular, when receiving signals transmitted using a transmission method that involves a change in phase applied to precoded (or precoded and switched) signals, data transmission effectiveness as well as signal quality are both improved in the LOS environment. 
     Although the present Embodiment is described as a reception device compatible with the transmission method described in Embodiment E1, and therefore having two antennas, no limitation is intended in this regard. The reception device may also have three or more antennas. In such cases, the data reception quality may be further improved by enhancing the diversity gain. Also, the transmission device of the broadcaster may have three or more transmit antennas and transmit three or more modulated signals. The same effects are achievable by accordingly increasing the number of antennas on the reception device of the terminal. Alternatively, the reception device may have only one antenna and apply maximum likelihood detection or approximate maximum likelihood detection. In such circumstances, the transmission method is preferably one that involves a change in phase of precoded (or precoded and switched) signals. 
     The transmission method need not be limited to the specific methods explained in the present description. As long as precoding occurs and is preceded or followed by a change in phase, the same results are obtainable for the present Embodiment. 
     Embodiment E3 
     The system of Embodiment E1, which applies, to the DVB-T2 standard, a transmission method involving a change in phase performed on precoded (or precoded and switched) signals, includes control information indicating the pilot insertion method in the L1 pre-signalling information. The present Embodiment describes a method of applying a transmission method that involves a change in phase performed on precoded signals (or precoded signals having switched basebands) when the pilot insertion method in the L1 pre-signalling information is changed. 
       FIGS.  89 A,  89 B,  90 A, and  90 B  illustrate sample frame configurations conforming to the DVB-T2 standard in the time-frequency domain in which a common frequency region is used in a transmission method by which a plurality of modulated signals are transmitted from a plurality of antennas. Here, the horizontal axes represent frequency, i.e., the carrier numbers, while the vertical axes represent time.  FIGS.  89 A and  90 A  illustrate frame configurations for modulated signal z 1  while  FIGS.  89 B and  90 B  illustrate frame configurations for modulated signal z 2 , both of which are as explained in the above Embodiments. The carrier numbers are labelled f 0 , f 1 , f 2 , and so on, while time is labelled t 1 , t 2 , t 3  and so on. Also, symbols indicated at the same carrier and time are simultaneous symbols at a common frequency. 
       FIGS.  89 A,  89 B,  90 A, and  90 B  illustrate examples of pilot symbol insertion positions conforming to the DVB-T2 standard. (In DVB-T2, eight methods of pilot insertion are possible when a plurality of antennas are used to transmit a plurality of modulated signals. Two of these are presently illustrated.) Two types of symbols are indicated, namely pilot symbols and data symbols. As described for other Embodiments, when the transmission method involves performing a change of phase on precoded signals (or precoded signals having switched basebands), or involves precoding using a fixed precoding matrix, then the data symbols of modulated signal z 1  are symbols of stream s 1  and stream s 2  that have undergone weighting, as are the data symbols of modulated signal z 2 . (However, a change in phase is also performed when the transmission scheme involves doing so) When space-time block codes or a spatial multiplexing MIMO system are used, the data symbols of modulated signal z 1  are the symbols of either stream s 1  or of stream s 2 , as are the symbols of modulated signal z 2 . In  FIGS.  89 A,  89 B,  90 A, and  90 B , the pilot symbols are labelled with an index, which is either PP1 or PP2. These represent pilot symbols using different configuration methods. As described above, eight methods of pilot insertion are possible in DVB-T2 (varying in terms of the frequency at which pilot symbols are inserted in the frame), one of which is indicated by the broadcaster.  FIGS.  89 A,  89 B,  90 A, and  90 B  illustrate two pilot insertion methods among these eight. As described in Embodiment E1, information pertaining to the pilot insertion method selected by the broadcaster is transmitted to the receiving terminal as the L1 pre-signalling data in the P2 symbol. 
     The following describes a method of applying a transmission method involving a change in phase performed on precoded signals (or precoded signals having switched basebands) complementing the pilot insertion method. In this example, the transmission method involves preparing ten different phase changing values, namely F[ 0 ], F[ 1 ], F[ 2 ], F[ 3 ], F[ 4 ], F[ 5 ], F[ 6 ], F[ 7 ], F[ 8 ], and F[ 9 ].  FIGS.  91 A and  91 B  illustrate the allocation of these phase changing values in the time-frequency domain frame configuration of  FIGS.  89 A and  89 B  when a transmission method involving a change in phase performed on precoded (or precoded and switched) signals is applied. Similarly,  FIGS.  92 A and  92 B  illustrate the allocation of these phase changing values in the time-frequency domain frame configuration of  FIGS.  90 A and  90 B  when a transmission method involving a change in phase performed on precoded (or precoded and switched) signals is applied. For example,  FIG.  91 A  illustrates the frame configuration of modulated signal z 1  while  FIG.  91 B  illustrates the frame configuration of modulated signal z 2 . In both cases, symbol # 1  at f 1 , t 1  is a symbol on which frequency modification has been performed using phase changing value F[ 1 ]. Accordingly, in  FIGS.  91 A,  91 B,  92 A , and  92 B, a symbol at carrier fx (where x=0, 1, 2, and so on), time ty (where y=1, 2, 3, and so on) is labelled #Z to indicate that frequency modification has been performed using phase changing value F[Z] on the symbol fx, ty. 
     Naturally, the insertion method (insertion interval) for the frequency-time frame configuration of  FIGS.  91 A and  91 B  differs from that of  FIGS.  92 A and  92 B . The transmission method in which a change of phase is performed on precoded signals (or precoded signals having switched basebands) is not applied to the pilot symbols. Therefore, although the same transmission method involving a change in phase performed on the same synchronized precoded (or precoded and switched) signals (for which a different number of phase changing values may have been prepared), the phase changing value assigned to a single symbol at a given carrier and time in  FIGS.  91 A and  91 B  may be different in  FIGS.  92 A and  92 B . This is made clear by reference to the drawings. For example, the symbol at f 5 , t 2  in  FIGS.  91 A and  91 B  is labelled # 7 , indicating that a change in phase has been performed thereon using phase changing value F[ 7 ]. On the other hand, the symbol at f 5 , t 2  in  FIGS.  92 A and  92 B  is labelled # 8 , indicating that a change in phase has been performed thereon using phase changing value F[ 8 ]. 
     Accordingly, although the broadcaster transmits control information indicating the pilot pattern (pilot insertion method) in the L1 pre-signalling information, when the transmission method selected by the broadcaster method involves a change in phase performed on precoded signals (or precoded signals having switched basebands), the control information may additionally indicate the phase changing value allocation method used in the selected method through the control information given by Table 3 or Table 4. Thus, the reception device of the terminal receiving the modulated signals transmitted by the broadcaster is able to determine the phase changing value allocation method by obtaining the control information indicating the pilot pattern in the L1 pre-signalling data. (This presumes that the transmission method selected by the broadcaster for PLP transmission from Table 3 or Table 4 is one that involves a change in phase on precoded signals (or precoded signals having switched basebands)). Although the above description uses the example of L1 pre-signalling data, the above-described control information may also be included in the first and second signalling data when, as described for  FIG.  83   , no P2 symbols are used. 
     The following describes further variant examples. Table 6 lists sample phase changing patterns and corresponding modulation schemes. 
     
       
         
           
               
               
               
             
               
                 TABLE 6 
               
               
                   
               
               
                 No. of  
                   
                 Phase 
               
               
                 Modulated  
                   
                 Changing  
               
               
                 Signals 
                 Modulation Scheme 
                 Pattern 
               
               
                   
               
             
            
               
                 2 
                 #1: QPSK, #2: QPSK 
                 #1: —, #2: A 
               
               
                 2 
                 #1: QPSK, #2: 16-QAM 
                 #1: —, #2: B 
               
               
                 2 
                 #1: 16-QAM, #2: 16-QAM 
                 #1: —, #2: C 
               
               
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
               
               
                   
               
            
           
         
       
     
     For example, as shown in Table 6, when the modulation scheme is indicated and the phase changing values to be used in the transmission method involving a change in phase performed on precoded signals (or precoded signals having switched basebands) have been determined, the above-described principles apply. That is, transmitting only the control information pertaining to the pilot pattern, the PLP transmission method, and the modulation scheme suffices to enable the reception device of the terminal to estimate the phase changing value allocation method (in the time-frequency domain) by obtaining this control information. In Table 6, the Phase Changing Method column lists a dash to indicate that no change in phase is performed, and lists #A, #B, or #C to indicate phase changing methods #A, #B, and #C. Similarly, as shown in Table 1, when the modulation scheme and the error-correcting coding method are indicated and the phase changing values to be used in the transmission method involving a change in phase of precoded signals (or precoded signals having switched basebands) have been determined, then transmitting only the control information pertaining to the pilot pattern, the PLP transmission method, the modulation scheme, and the error-correcting codes in the P2 symbol suffices to enable the reception device of the terminal to estimate the phase changing value allocation method (in the time-frequency domain) by obtaining this control information. 
     However, unlike Table 1 and Table 6, two or more different types of transmission scheme involving a change in phase performed on precoded signals (or precoded signals having switched basebands) may be selected, despite the modulation scheme having been determined (For example, the transmission schemes may have a different period (cycle), or use different phase changing values). Alternatively, two or more different types of transmission scheme involving a change in phase performed on precoded signals (or precoded signals having switched basebands) may be selected, despite the modulation scheme and the error-correction scheme having been determined. Furthermore, two or more different types of transmission scheme involving a change in phase performed on precoded signals (or precoded signals having switched basebands) may be selected, despite the error-correction scheme having been determined. In such cases, as shown in Table 4, the transmission scheme involves switching between phase changing values. However, information pertaining to the allocation scheme of the phase changing values (in the time-frequency domain) may also be transmitted. 
     Table 7 lists control information configuration examples for information pertaining to such allocation methods. 
     
       
         
           
               
               
             
               
                 TABLE 7 
               
               
                   
               
               
                 PHASE_FRAME_ARRANGEMENT  
                   
               
               
                 (2-bit) 
                 Control Information 
               
               
                   
               
             
            
               
                 00 
                 allocation scheme #1 
               
               
                 01 
                 allocation scheme #2 
               
               
                 10 
                 allocation scheme #3 
               
               
                 11 
                 allocation scheme #4 
               
               
                   
               
            
           
         
       
     
     For example, suppose that the transmission device of the broadcaster selects  FIGS.  89 A and  89 B  as the pilot pattern insertion method, and selects transmission method A, which involves a change in phase on precoded signals (or precoded signals having switched basebands). Thus, the transmission device may select  FIGS.  91 A and  91 B  or  FIGS.  93 A and  93 B  as the phase changing value allocation method (in the time-frequency domain). For example, when the transmission device selects  FIGS.  91 A and  91 B , the PHASE FRAME ARRANGEMENT information of Table 7 is set to 00. When the transmission device selects  FIGS.  93 A and  93 B , the PHASE FRAME ARRANGEMENT information is set to 01. As such, the reception device is able to determine the phase changing value allocation method (in the time-frequency domain) by obtaining the control information of Table 7. The control information of Table 7 is also applicable to transmission by the P2 symbol, and to transmission by the first and second signalling data. 
     As described above, a phase changing value allocation method for the transmission method involving a change in phase performed on precoded (or precoded and switched) signals may be realized through the pilot insertion method. In addition, by reliably transmitting such allocation method information to the receiving party, the reception device derives the dual benefits of improved data transmission efficiency and enhanced received signal quality. 
     Although the present Embodiment describes a broadcaster using two transmit signals, the same applies to broadcasters using a transmission device having three or more transmit antennas transmitting three or more signals. The transmission method need not be limited to the specific methods explained in the present description. As long as precoding occurs and is preceded or followed by a change in phase, the same results are obtainable for the present Embodiment. 
     The pilot signal configuration method is not limited to the present Embodiment. When the transmission method involves performing a change of phase on precoded (or precoded and switched) signals, the reception device need only implement the relationship given by Math. 48 (formula 48) (e.g., the reception device may know the pilot pattern signals transmitted by the transmission device in advance). This applies to all Embodiments discussed in the present description. 
     The transmission devices pertaining to the present invention, as illustrated by  FIGS.  3 ,  4 ,  12 ,  13 ,  51 ,  52 ,  67 ,  70 ,  76 ,  85   , and so on transmit two modulated signals, namely modulated signal # 1  and modulated signal # 2 , on two different transmit antennas. The average transmission power of the modulated signals # 1  and # 2  may be set freely. For example, when the two modulated signals each have a different average transmission power, conventional transmission power control technology used in wireless transmission systems may be applied thereto. Therefore, the average transmission power of modulated signals # 1  and # 2  may differ. In such circumstances, transmission power control may be applied to the baseband signals (e.g., when mapping is performed using the modulation scheme), or may be performed by a power amplifier immediately before the antenna. 
     Embodiment F1 
     The schemes for regularly performing phase change on the modulated signal after precoding described in Embodiments 1 through 4, Embodiment A1, Embodiments C1 through C7, Embodiments D1 through D3 and Embodiments E1 through E3 are applicable to any baseband signals s 1  and s 2  mapped in the I-Q plane. Therefore, in Embodiments 1 through 4, Embodiment A1, Embodiments C1 through C7, Embodiments D1 through D3 and Embodiments E1 through E3, the baseband signals s 1  and s 2  have not been described in detail. On the other hand, when the scheme for regularly performing phase change on the modulated signal after precoding is applied to the baseband signals s 1  and s 2  generated from the error correction coded data, excellent reception quality can be achieved by controlling average power (average value) of the baseband signals s 1  and s 2 . In the present embodiment, the following describes a scheme of setting the average power of s 1  and s 2  when the scheme for regularly performing phase change on the modulated signal after precoding is applied to the baseband signals s 1  and s 2  generated from the error correction coded data. 
     As an example, the modulation schemes for the baseband signal s 1  and the baseband signal s 2  are described as QPSK and 16QAM, respectively. 
     Since the modulation scheme for s 1  is QPSK, s 1  transmits two bits per symbol. Let the two bits to be transmitted be referred to as b 0  and b 1 . On the other hand, since the modulation scheme for s 2  is 16QAM, s 2  transmits four bits per symbol. Let the four bits to be transmitted be referred to as b 2 , b 3 , b 4  and b 5 . The transmission device transmits one slot composed of one symbol for s 1  and one symbol for s 2 , i.e. six bits b 0 , b 1 , b 2 , b 3 , b 4  and b 5  per slot. 
     For example, in  FIG.  95    as an example of signal point layout in the I-Q plane for 16QAM, (b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 0) is mapped onto (I, Q)=(3×g, 3×g), (b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 1) is mapped onto (I, Q)=(3×g, 1×g), (b 2 , b 3 , b 4 , b 5 )=(0, 0, 1, 0) is mapped onto (I, Q)=(1×g, 3×g), (b 2 , b 3 , b 4 , b 5 )=(0, 0, 1, 1) is mapped onto (I, Q)=(1×g, 1×g), (b 2 , b 3 , b 4 , b 5 )=(0, 1, 0, 0) is mapped onto (I, Q)=(3×g, −3×g), (b 2 , b 3 , b 4 , b 5 )=(1, 1, 1, 0) is mapped onto (I, Q)=(−1×g, −3×g), and (b 2 , b 3 , b 4 , b 5 )=(1, 1, 1, 1) is mapped onto (I, Q)=(−1×g, −1×g). Note that b 2  through b 5  shown on the top right of  FIG.  95    shows the bits and the arrangement of the numbers shown on the I-Q plane. 
     Also, in  FIG.  96    as an example of signal point layout in the I-Q plane for QPSK, (b 0 , b 1 )=(0, 0) is mapped onto (I, Q)=(1×h, 1×h), (b 0 , b 1 )=(0, 1) is mapped onto (I, Q)=(1×h, −1×h), (b 0 , b 1 )=(1, 0) is mapped onto (I, Q)=(−1×h, 1×h), and (b 0 , b 1 )=(1, 1) is mapped onto (I, Q)=(−1×h, −1×h). Note that b 0  and b 1  shown on the top right of  FIG.  96    shows the bits and the arrangement of the numbers shown on the I-Q plane. 
     Here, assume that the average power of s 1  is equal to the average power of s 2 , i.e. h shown in  FIG.  96    is represented by formula 78 and g shown in  FIG.  95    is represented by formula 79. 
     
       
         
           
             
               
                 
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                     . 
                     
                         
                     
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       FIG.  97    shows the log-likelihood ratio obtained by the reception device in this case.  FIG.  97    schematically shows absolute values of the log-likelihood ratio for b 0  through b 5  described above when the reception device obtains the log-likelihood ratio. In  FIG.  97 ,  9700    is the absolute value of the log-likelihood ratio for b 0 ,  9701  is the absolute value of the log-likelihood ratio for b 1 ,  9702  is the absolute value of the log-likelihood ratio for b 2 ,  9703  is the absolute value of the log-likelihood ratio for b 3 ,  9704  is the absolute value of the log-likelihood ratio for b 4 , and  9705  is the absolute value of the log-likelihood ratio for b 5 . In this case, as shown in  FIG.  97   , when the absolute values of the log-likelihood ratio for b 0  and b 1  transmitted in QPSK are compared with the absolute values of the log-likelihood ratio for b 2  through b 5  transmitted in 16QAM, the absolute values of the log-likelihood ratio for b 0  and b 1  are higher than the absolute values of the log-likelihood ratio for b 2  through b 5 . That is, reliability of b 0  and b 1  in the reception device is higher than the reliability of b 2  through b 5  in the reception device. This is because of the following reason. When h is represented by formula 79 in  FIG.  95   , a minimum Euclidian distance between signal points in the I-Q plane for QPSK is as follows.
 
[Math. 80]
 
√{square root over (2 z )}  (formula 80)
 
     On the other hand, when h is represented by formula 78 in  FIG.  78   , a minimum Euclidian distance between signal points in the I-Q plane for 16QAM is as follows. 
     
       
         
           
             
               
                 
                   [ 
                   
                     Math 
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                     80 
                   
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     This is the reason. 
     If the reception device performs error correction decoding (e.g. belief propagation decoding such as a sum-product decoding in a case where the communication system uses LDPC codes) under this situation, due to a difference in reliability that “the absolute values of the log-likelihood ratio for b 0  and b 1  are higher than the absolute values of the log-likelihood ratio for b 2  through b 5 ”, a problem that the data reception quality degrades in the reception device by being affected by the absolute values of the log-likelihood ratio for b 2  through b 5  arises. 
     In order to overcome the problem, the difference between the absolute values of the log-likelihood ratio for b 0  and b 1  and the absolute values of the log-likelihood ratio for b 2  through b 5  should be reduced compared with  FIG.  97   , as shown in  FIG.  98   . 
     Therefore, it is considered that the average power (average value) of s 1  is made to be different from the average power (average value) of s 2 .  FIGS.  99  and  100    each show an example of the structure of the signal processor relating to a power changer (although being referred to as the power changer here, the power changer may be referred to as an amplitude changer or a weight unit) and the weighting (precoding) unit. In  FIG.  99   , elements that operate in a similar way to  FIG.  3    and  FIG.  6    bear the same reference signs. Also, in  FIG.  100   , elements that operate in a similar way to  FIG.  3   ,  FIG.  6    and  FIG.  99    bear the same reference signs. 
     The following explains some examples of operations of the power changer. 
     Example 1 
     First, an example of the operation is described using  FIG.  99   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme QPSK. The mapping scheme for s 1 ( t ) is as shown in  FIG.  96   , and h is as represented by formula 78. Also, let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is as represented by formula 79. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 16QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 16QAM by u. Let u be a real number, and u&gt;1.0. 
     Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , the following formula is satisfied. 
     
       
         
           
             
               
                 
                   
                       
                   
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     Therefore, a ratio of the average power for QPSK to the average power for 16QAM is set to 1:u 2 . With this structure, the reception device is in a reception condition in which the absolute value of the log-likelihood ratio shown in  FIG.  98    is obtained. Therefore, data reception quality is improved in the reception device. 
     The following describes a case where u in the ratio of the average power for QPSK to the average power for 16QAM 1:u 2  is set as shown in the following formula.
 
[Math. 83]
 
 u =√{square root over (5)}(formula 83)=
 
     In this case, the minimum Euclidian distance between signal points in the I-Q plane for QPSK and the minimum Euclidian distance between signal points in the I-Q plane for 16QAM can be the same. Therefore, excellent reception quality can be achieved. 
     The condition that the minimum Euclidian distances between signal points in the I-Q plane for two different modulation schemes are equalized, however, is a mere example of the scheme of setting the ratio of the average power for QPSK to the average power for 16QAM. For example, according to other conditions such as a code length and a coding rate of an error correction code used for error correction codes, excellent reception quality may be achieved when the value u for power change is set to a value (higher value or lower value) different from the value at which the minimum Euclidian distances between signal points in the I-Q plane for two different modulation schemes are equalized. Considering the processing efficiency, a scheme of setting the value u as shown in the following formula is considered, for example.
 
[Math. 84]
 
 u =√{square root over (2)}  (formula 84)
 
     This will be described later in detail. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the value u for power change is set based on the control signal ( 9900 ). The following describes setting of the value u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 1-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction coding used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, a value for power change set according to a block length X is referred to as u LX    
     For example, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . In this case, for example, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied. What is important is that two or more values exist in u L1000 , u L1500  and u L3000 ). 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change. 
     Example 1-2 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, a value for power change set according to a coding rate rx is referred to as u rX . 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . In this case, for example, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u r1 =u r2  may be satisfied. What is important is that two or more values exist in u r1 , u r2  and u r3 ). 
     Note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change. 
     Example 1-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to QPSK and the modulation scheme for s 2  is changed from 16QAM to 64QAM by the control signal (or can be set to either 16QAM or 64QAM) is considered. Note that, in a case where the modulation scheme for s 2 ( t ) is 64QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  101   . In  FIG.  101   , k is represented by the following formula. 
     
       
         
           
             
               
                 
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     By performing mapping in this way, the average power obtained when h in  FIG.  96    for QPSK is represented by formula 78 becomes equal to the average power obtained when g in  FIG.  95    for 16QAM is represented by formula 79. In the mapping in 64QAM, the values I and Q are determined from an input of six bits. In this regard, the mapping 64QAM may be performed similarly to the mapping in QPSK and 16QAM. 
     That is to say, in  FIG.  101    as an example of signal point layout in the I-Q plane for 64QAM, (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 0, 0, 0) is mapped onto (I, Q)=(7×k, 7×k), (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 0, 0, 1) is mapped onto (I, Q)=(7×k, 5×k), (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 0, 1, 0) is mapped onto (I, Q)=(5×k, 7×k), (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 0, 1, 1) is mapped onto (I, Q)=(5×k, 5×k), (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(0, 0, 0, 1, 0, 0) is mapped onto (I, Q)=(7×k, 1×k), (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(1, 1, 1, 1, 1, 0) is mapped onto (I, Q)=(−3×k, −1×k), and (b 0 , b 1 , b 2 , b 3 , b 4 , b 5 )=(1, 1, 1, 1, 1, 1) is mapped onto (I, Q)=(−3×k, −3×k). Note that b 0  through b 5  shown on the top right of  FIG.  101    shows the bits and the arrangement of the numbers shown on the I-Q plane. 
     In  FIG.  99   , the power changer  9901 B sets such that u=u 16  when the modulation scheme for s 2  is 16QAM, and sets such that u=u 64  when the modulation scheme for s 2  is 64QAM. In this case, due to the relationship between minimum Euclidian distances, by setting such that u 16 &lt;u 64 , excellent data reception quality is obtained in the reception device when the modulation scheme for s 2  is either 16QAM or 64QAM. 
     Note that, in the above description, the “modulation scheme for s 1  is fixed to QPSK”. It is also considered that the modulation scheme for s 2  is fixed to QPSK. In this case, power change is assumed to be not performed for the fixed modulation scheme (here, QPSK), and to be performed for a plurality of modulation schemes that can be set (here, 16QAM and 64QAM). That is to say, in this case, the transmission device does not have the structure shown in  FIG.  99   , but has a structure in which the power changer  9901 B is eliminated from the structure in  FIG.  99    and a power changer is provided to a s 1 ( t )-side. When the fixed modulation scheme (here, QPSK) is set to s 2 , the following formula 86 is satisfied. 
     
       
         
           
             
               
                 
                   
                       
                   
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     When the modulation scheme for s 2  is fixed to QPSK and the modulation scheme for s 1  is changed from 16QAM to 64QAM (is set to either 16QAM or 64QAM), the relationship u 16 &lt;u 64  should be satisfied (note that a multiplied value for power change in 16QAM is u 16 , a multiplied value for power change in 64QAM is u 64 , and power change is not performed in QPSK). 
     Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of QPSK and 16QAM, a set of 16QAM and QPSK, a set of QPSK and 64QAM and a set of 64QAM and QPSK, the relationship u 16 &lt;u 64  should be satisfied. 
     The following describes a case where the above-mentioned description is generalized. 
     Let the modulation scheme for s 1  be fixed to a modulation scheme C in which the number of signal points in the I-Q plane is c. Also, let the modulation scheme for s 2  be set to either a modulation scheme A in which the number of signal points in the I-Q plane is a or a modulation scheme B in which the number of signal points in the I-Q plane is b (a&gt;b&gt;c) (however, let the average power (average value) for s 2  in the modulation scheme A be equal to the average power (average value) for s 2  in the modulation scheme B). 
     In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s 2  is u a . Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s 2  is u b . In this case, when the relationship u b &lt;u a  is satisfied, excellent data reception quality is obtained in the reception device. 
     Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship u b &lt;u a  should be satisfied. Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship u b &lt;u a  should be satisfied. 
     Example 2 
     The following describes an example of the operation different from that described in Example 1, using  FIG.  99   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme 64QAM. The mapping scheme for s 1 ( t ) is as shown in  FIG.  101   , and k is as represented by formula 85. Also, let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is as represented by formula 79. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 16QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 16QAM by u. Let u be a real number, and u&lt;1.0. Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , the following formula is satisfied. 
     Therefore, a ratio of the average power for 64QAM to the average power for 16QAM is set to 1:u 2 . With this structure, the reception device is in a reception condition as shown in  FIG.  98   . Therefore, data reception quality is improved in the reception device. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the value u for power change is set based on the control signal ( 9900 ). The following describes setting of the value u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 2-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, a value for power change set according to a block length X is referred to as u LX    
     For example, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . In this case, for example, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied. What is important is that two or more values exist in u L1000 , u L1500  and u L3000 ). 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change. 
     Example 2-2 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, a value for power change set according to a coding rate rx is referred to as u rx . 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . In this case, for example, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u r1 =u r2  may be satisfied. What is important is that two or more values exist in u r1 , u r2  and u r3 ). Note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change. 
     Example 2-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to 64QAM and the modulation scheme for s 2  is changed from 16QAM to QPSK by the control signal (or can be set to either 16QAM or QPSK) is considered. In a case where the modulation scheme for s 1  is 64QAM, the mapping scheme for s 1 ( t ) is as shown in  FIG.  101   , and k is represented by formula 85 in  FIG.  101   . In a case where the modulation scheme for s 2  is 16QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is represented by formula 79 in  FIG.  95   . Also, in a case where the modulation scheme for s 2 ( t ) is QPSK, the mapping scheme for s 2 ( t ) is as shown in  FIG.  96   , and h is represented by formula 78 in  FIG.  96   . 
     By performing mapping in this way, the average power in 16QAM becomes equal to the average power (average value) in QPSK. 
     In  FIG.  99   , the power changer  9901 B sets such that u=u 16  when the modulation scheme for s 2  is 16QAM, and sets such that u=u 4  when the modulation scheme for s 2  is QPSK. In this case, due to the relationship between minimum Euclidian distances, by setting such that u 4 &lt;u 16 , excellent data reception quality is obtained in the reception device when the modulation scheme for s 2  is either 16QAM or QPSK. 
     Note that, in the above description, the modulation scheme for s 1  is fixed to 64QAM. When the modulation scheme for s 2  is fixed to 64QAM and the modulation scheme for s 1  is changed from 16QAM to QPSK (is set to either 16QAM or QPSK), the relationship u 4 &lt;u 16  should be satisfied (the same considerations should be made as the example 1-3) (note that a multiplied value for power change in 16QAM is u 16 , a multiplied value for power change in QPSK is u 4 , and power change is not performed in 64QAM). Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of 64QAM and 16QAM, a set of 16QAM and 64QAM, a set of 64QAM and QPSK and a set of QPSK and 64QAM, the relationship u 4 &lt;u 16  should be satisfied. 
     The following describes a case where the above-mentioned description is generalized. 
     Let the modulation scheme for s 1  be fixed to a modulation scheme C in which the number of signal points in the I-Q plane is c. Also, let the modulation scheme for s 2  be set to either a modulation scheme A in which the number of signal points in the I-Q plane is a or a modulation scheme B in which the number of signal points in the I-Q plane is b (c&gt;b&gt;a) (however, let the average power (average value) for s 2  in the modulation scheme A be equal to the average power (average value) for s 2  in the modulation scheme B). 
     In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s 2  is u a . Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s 2  is u b . In this case, when the relationship u a &lt;u b  is satisfied, excellent data reception quality is obtained in the reception device. 
     Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship u a &lt;u b  should be satisfied. Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship u a &lt;u b  should be satisfied. 
     Example 3 
     The following describes an example of the operation different from that described in Example 1, using  FIG.  99   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 1 ( t ) is as shown in  FIG.  95   , and g is as represented by formula 79. Let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 64QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  101   , and k is as represented by formula 85. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 64QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 64QAM by u. Let u be a real number, and u&gt;1.0. Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , the following formula is satisfied. 
     Therefore, a ratio of the average power for 16QAM to the average power for 64QAM is set to 1:u 2 . With this structure, the reception device is in a reception condition as shown in  FIG.  98   . Therefore, data reception quality is improved in the reception device. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the value u for power change is set based on the control signal ( 9900 ). The following describes setting of the value u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 3-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, a value for power change set according to a block length X is referred to as u LX    
     For example, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . In this case, for example, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied. What is important is that two or more values exist in u L1000 , u L1500  and u L3000 ). 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change. 
     Example 3-2 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changer ( 9901 B) sets the value u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, a value for power change set according to a coding rate rx is referred to as u rx . 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . In this case, for example, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u r1 =u r2  may be satisfied. What is important is that two or more values exist in u r1 , u r2  and u r3 ). 
     Note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change. 
     Example 3-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to 16QAM and the modulation scheme for s 2  is changed from 64QAM to QPSK by the control signal (or can be set to either 64QAM or QPSK) is considered. In a case where the modulation scheme for s 1  is 16QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is represented by formula 79 in  FIG.  95   . In a case where the modulation scheme for s 2  is 64QAM, the mapping scheme for s 1 ( t ) is as shown in  FIG.  101   , and k is represented by formula 85 in  FIG.  101   . Also, in a case where the modulation scheme for s 2 ( t ) is QPSK, the mapping scheme for s 2 ( t ) is as shown in  FIG.  96   , and h is represented by formula 78 in  FIG.  96   . 
     By performing mapping in this way, the average power in 16QAM becomes equal to the average power in QPSK. 
     In  FIG.  99   , the power changer  9901 B sets such that u=u 64  when the modulation scheme for s 2  is 64QAM, and sets such that u=u 4  when the modulation scheme for s 2  is QPSK. In this case, due to the relationship between minimum Euclidian distances, by setting such that u 4 &lt;u 64 , excellent data reception quality is obtained in the reception device when the modulation scheme for s 2  is either 16QAM or 64QAM. 
     Note that, in the above description, the modulation scheme for s 1  is fixed to 16QAM. When the modulation scheme for s 2  is fixed to 16QAM and the modulation scheme for s 1  is changed from 64QAM to QPSK (is set to either 64QAM or QPSK), the relationship u 4 &lt;u 64  should be satisfied (the same considerations should be made as the example 1-3) (note that a multiplied value for power change in 64QAM is u 64 , a multiplied value for power change in QPSK is u 4 , and power change is not performed in 16QAM). Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of 16QAM and 64QAM, a set of 64QAM and 16QAM, a set of 16QAM and QPSK and a set of QPSK and 16QAM, the relationship u 4 &lt;u 64  should be satisfied. 
     The following describes a case where the above-mentioned description is generalized. 
     Let the modulation scheme for s 1  be fixed to a modulation scheme C in which the number of signal points in the I-Q plane is c. Also, let the modulation scheme for s 2  be set to either a modulation scheme A in which the number of signal points in the I-Q plane is a or a modulation scheme B in which the number of signal points in the I-Q plane is b (c&gt;b&gt;a) (however, let the average power (average value) for s 2  in the modulation scheme A be equal to the average power (average value) for s 2  in the modulation scheme B). 
     In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s 2  is u a . Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s 2  is u b . In this case, when the relationship u a &lt;u b  is satisfied, excellent data reception quality is obtained in the reception device. 
     Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship u a &lt;u b  should be satisfied. Also, when a set of the modulation scheme for s 1  and the modulation scheme for s 2  can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship u a &lt;u b  should be satisfied. 
     Example 4 
     The case where power change is performed for one of the modulation schemes for s 1  and s 2  has been described above. The following describes a case where power change is performed for both of the modulation schemes for s 1  and s 2 . 
     An example of the operation is described using  FIG.  100   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme QPSK. The mapping scheme for s 1 ( t ) is as shown in  FIG.  96   , and h is as represented by formula 78. Also, let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is as represented by formula 79. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 A) receives a (mapped) baseband signal  307 A for the modulation scheme QPSK and the control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be v, the power changer outputs a signal ( 9902 A) obtained by multiplying the (mapped) baseband signal  307 A for the modulation scheme QPSK by v. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 16QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 16QAM by u. Then, let u=v×w (w&gt;1.0). 
     Letting the precoding matrix used in the scheme for regularly performing phase change be F[t], formula 87 shown next is satisfied. 
     Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , formula 87 shown next is satisfied. 
     
       
         
           
             
               
                 
                   
                       
                   
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     Therefore, a ratio of the average power for QPSK to the average power for 16QAM is set to v 2 :u 2 =v 2 :v 2 ×w 2 =1:w 2 . With this structure, the reception device is in a reception condition as shown in  FIG.  98   . Therefore, data reception quality is improved in the reception device. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the values v and u for power change are set based on the control signal ( 9900 ). The following describes setting of the values v and u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 4-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, values for power change set according to the block length X are referred to as v LX  and u LX . 
     For example, when 1000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1000 . When 1500 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1500 . When 3000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to V L3000 . 
     On the other hand, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . 
     In this case, for example, by setting v L1000 , v L1500  and v L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied, and v L1000 =v L1500  may be satisfied. What is important is that two or more values exist in a set of v L1000 , v L1500  and v L3000 , and that two or more values exist in a set of u L1000 , u L1500  and u L3000 ). Note that, as described above, v LX  and u LX  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values u LX  for power change when the code length is set, and performs power change. Another important point is that two or more values v LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values v LX  for power change when the code length is set, and performs power change. 
     Example 4-2 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 A) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, values for power change set according to the coding rate rx are referred to as v rx  and u rx . 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r1 . When r 2  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r2 . When r 3  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r3 . 
     Also, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . 
     In this case, for example, by setting v r1 , v r2  and v r3  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, v r1 =v r2  may be satisfied, and u r1 =u r2  may be satisfied. What is important is that two or more values exist in a set of v r1 , v r2  and v r3 , and that two or more values exist in a set of u r1 , u r2  and u r3 ). Note that, as described above, v rX  and u rX  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Also, note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u rx  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values up, for power change when the coding rate is set, and performs power change. Another important point is that two or more values v rx  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values v rx  for power change when the coding rate is set, and performs power change. 
     Example 4-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to QPSK and the modulation scheme for s 2  is changed from 16QAM to 64QAM by the control signal (or can be set to either 16QAM or 64QAM) is considered. In a case where the modulation scheme for s 1  is QPSK, the mapping scheme for s 1 ( t ) is as shown in  FIG.  96   , and h is represented by formula 78 in  FIG.  96   . In a case where the modulation scheme for s 2  is 16QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is represented by formula 79 in  FIG.  95   . Also, in a case where the modulation scheme for s 2 ( t ) is 64QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  101   , and k is represented by formula 85 in  FIG.  101   . 
     In  FIG.  100   , when the modulation scheme for s 1  is QPSK and the modulation scheme for s 2  is 16QAM, assume that v=α and u=α×w 16 . In this case, the ratio between the average power of QPSK and the average power of 16QAM is v 2 :u 2 =α 2 :α 2 ×w 16   2 =1:w 16   2 . 
     In  FIG.  100   , when the modulation scheme for s 1  is QPSK and the modulation scheme for s 2  is 64QAM, assume that v=β and u=β×w 64 . In this case, the ratio between the average power of QPSK and the average power of 64QAM is v:u=β 2 :β 2 ×w 64   2 =1:w 64   2 . In this case, according to the minimum Euclidean distance relationship, the reception device achieves high data reception quality when 1.0&lt;w 16 &lt;w 64 , regardless of whether the modulation scheme for s 2  is 16QAM or 64QAM. 
     Note that although “the modulation scheme for s 1  is fixed to QPSK” in the description above, it is possible that “the modulation scheme for s 2  is fixed to QPSK”. In this case, power change is assumed to be not performed for the fixed modulation scheme (here, QPSK), and to be performed for a plurality of modulation schemes that can be set (here, 16QAM and 64QAM). When the fixed modulation scheme (here, QPSK) is set to s 2 , the following formula 88 is satisfied. 
     
       
         
           
             
               
                 
                   
                       
                   
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     Given that, even when “the modulation scheme for s 2  is fixed to QPSK and the modulation scheme for s 1  is changed from 16QAM to 64QAM (set to either 16QAM or 64QAM)”, 1.0&lt;w 16 &lt;w 64  should be fulfilled. (Note that the value used for the multiplication for the power change in the case of 16QAM is u=α×w m , the value used for the multiplication for the power change in the case of 64QAM is u=β×w 64 , the value used for the power change in the case of QPSK is v=α when the selectable modulation scheme is 16QAM and v=β when the selectable modulation scheme is 64QAM.) Also, when the set of (the modulation scheme for s 1 , the modulation scheme for s 2 ) is selectable from the sets of (QPSK, 16QAM), (16QAM, QPSK), (QPSK, 64QAM) and (64QAM, QPSK), 1.0&lt;w 16 &lt;w 64  should be fulfilled. 
     The following describes a case where the above-mentioned description is generalized. 
     For generalization, assume that the modulation scheme for s 1  is fixed to a modulation scheme C with which the number of signal points in the I-Q plane is c. Also assume that the modulation scheme for s 2  is selectable from a modulation scheme A with which the number of signal points in the I-Q plane is a and a modulation scheme B with which the number of signal points in the I-Q plane is b (a&gt;b&gt;c). In this case, when the modulation scheme for s 2  is set to the modulation scheme A, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme A, is 1:w a   2 . Also, when the modulation scheme for s 2  is set to the modulation scheme B, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme B, is 1:w b   2 . If this is the case, the reception device achieves a high data reception quality when w b &lt;w a  is fulfilled. 
     Note that although “the modulation scheme for s 1  is fixed to C” in the description above, even when “the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (set to either the modulation scheme A or the modulation scheme B), the average powers should fulfill w b &lt;w a . (If this is the case, as with the description above, when the average power of the modulation scheme C is 1, the average power of the modulation scheme A is w a   2 , and the average power of the modulation scheme B is w b   2 .) Also, when the set of (the modulation scheme for s 1 , the modulation scheme for s 2 ) is selectable from the sets of (the modulation scheme C, the modulation scheme A), (the modulation scheme A, the modulation scheme C), (the modulation scheme C, the modulation scheme B) and (the modulation scheme B, the modulation scheme C), the average powers should fulfill w b &lt;w a . 
     Example 5 
     The following describes an example of the operation different from that described in Example 4, using  FIG.  100   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme 64QAM. The mapping scheme for s 1 ( t ) is as shown in  FIG.  86   , and k is as represented by formula 85. Also, let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is as represented by formula 79. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 A) receives a (mapped) baseband signal  307 A for the modulation scheme 64QAM and the control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be v, the power changer outputs a signal ( 9902 A) obtained by multiplying the (mapped) baseband signal  307 A for the modulation scheme 64QAM by v. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 16QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 16QAM by u. Then, let u=v×w (w&lt;1.0). 
     Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , formula 87 shown above is satisfied. 
     Therefore, a ratio of the average power for 64QAM to the average power for 16QAM is set to v 2 :u 2 =v 2 :v 2 ×w 2 =1:w 2 . With this structure, the reception device is in a reception condition as shown in  FIG.  98   . Therefore, data reception quality is improved in the reception device. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the values v and u for power change are set based on the control signal ( 9900 ). The following describes setting of the values v and u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 5-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, values for power change set according to the block length X are referred to as v LX  and u LX . 
     For example, when 1000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1000 . When 1500 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1500 . When 3000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L3000 . 
     On the other hand, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . 
     In this case, for example, by setting v L1000 , v L1500  and v L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied, and v L1000 =v L1000  may be satisfied. What is important is that two or more values exist in a set of v L1000 , v L1500  and v L3000 , and that two or more values exist in a set of u L1000 , u L1500  and u L3000 ). Note that, as described above, v LX  and u LX  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values u LX  for power change when the code length is set, and performs power change. Another important point is that two or more values v LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values v LX  for power change when the code length is set, and performs power change. 
     Example 5-2 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 A) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, values for power change set according to the coding rate rx are referred to as v rx  and up. 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r1 . When r 2  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r2 . When r 3  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r3 . 
     Also, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . 
     In this case, for example, by setting v r1 , v r2  and v r3  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, v r1 =v r2  may be satisfied, and u r1 =u r2  may be satisfied. What is important is that two or more values exist in a set of v r1 , v r2  and v r3 , and that two or more values exist in a set of u r1 , u r2  and u r3 ). Note that, as described above, v rX  and u rX  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Also, note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u rx  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values u rx  for power change when the coding rate is set, and performs power change. Another important point is that two or more values v rx  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values v rx  for power change when the coding rate is set, and performs power change. 
     Example 5-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to 64QAM and the modulation scheme for s 2  is changed from 16QAM to QPSK by the control signal (or can be set to either 16QAM or QPSK) is considered. In a case where the modulation scheme for s 1  is 64QAM, the mapping scheme for s 1 ( t ) is as shown in  FIG.  101   , and k is represented by formula 85 in  FIG.  101   . In a case where the modulation scheme for s 2  is 16QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  95   , and g is represented by formula 79 in  FIG.  95   . Also, in a case where the modulation scheme for s 2 ( t ) is QPSK, the mapping scheme for s 2 ( t ) is as shown in  FIG.  96   , and h is represented by formula 78 in  FIG.  96   . 
     In  FIG.  100   , when the modulation scheme for s 1  is 64QAM and the modulation scheme for s 2  is 16QAM, assume that v=a and u=α×w 16 . In this case, the ratio between the average power of 64QAM and the average power of 16QAM is v 2 :u 2 =α 2 :α 2 ×w 16   2 =1:w 16   2 . 
     In  FIG.  100   , when the modulation scheme for s 1  is 64QAM and the modulation scheme for s 2  is QPSK, assume that v=β and u=β×w 4 . In this case, the ratio between the average power of 64QAM and the average power of QPSK is v 2 :u 2 =β 2 :β 2 ×w 4   2 =1:w 4   2 . In this case, according to the minimum Euclidean distance relationship, the reception device achieves a high data reception quality when w 4 &lt;w 16 &lt;1.0, regardless of whether the modulation scheme for s 2  is 16QAM or QPSK. 
     Note that although “the modulation scheme for s 1  is fixed to 64QAM” in the description above, it is possible that “the modulation scheme for s 2  is fixed to 64QAM and the modulation scheme for s 1  is changed from 16QAM to QPSK (set to either 16QAM or QPSK)”, w 4 &lt;w 16 &lt;1.0 should be fulfilled. (The same as described in Example 4-3). (Note that the value used for the multiplication for the power change in the case of 16QAM is u=α×w 16 , the value used for the multiplication for the power change in the case of QPSK is u=β×w 4 , the value used for the power change in the case of 64QAM is v=α when the selectable modulation scheme is 16QAM and v=β when the selectable modulation scheme is QPSK). Also, when the set of (the modulation scheme for s 1 , the modulation scheme for s 2 ) is selectable from the sets of (64QAM, 16QAM), (16QAM, 64QAM), (64QAM, QPSK) and (QPSK, 64QAM), w 4 &lt;w 16 &lt;1.0 should be fulfilled. 
     The following describes a case where the above-mentioned description is generalized. 
     For generalization, assume that the modulation scheme for s 1  is fixed to a modulation scheme C with which the number of signal points in the I-Q plane is c. Also assume that the modulation scheme for s 2  is selectable from a modulation scheme A with which the number of signal points in the I-Q plane is a and a modulation scheme B with which the number of signal points in the I-Q plane is b (c&gt;b&gt;a). In this case, when the modulation scheme for s 2  is set to the modulation scheme A, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme A, is 1:w a   2 . Also, when the modulation scheme for s 2  is set to the modulation scheme B, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme B, is 1:w b   2 . If this is the case, the reception device achieves a high data reception quality when w a &lt;w b  is fulfilled. 
     Note that although “the modulation scheme for s 1  is fixed to C” in the description above, even when “the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (set to either the modulation scheme A or the modulation scheme B), the average powers should fulfill w a &lt;w b . (If this is the case, as with the description above, when the average power of the modulation scheme is C, the average power of the modulation scheme A is w a   g , and the average power of the modulation scheme B is w b   2 .) Also, when the set of (the modulation scheme for s 1 , the modulation scheme for s 2 ) is selectable from the sets of (the modulation scheme C, the modulation scheme A), (the modulation scheme A, the modulation scheme C), (the modulation scheme C, the modulation scheme B) and (the modulation scheme B, the modulation scheme C), the average powers should fulfill w a &lt;w b . 
     Example 6 
     The following describes an example of the operation different from that described in Example 4, using  FIG.  100   . Let s 1 ( t ) be the (mapped) baseband signal for the modulation scheme 16QAM. The mapping scheme for s 1 ( t ) is as shown in  FIG.  101   , and g is as represented by formula 79. Let s 2 ( t ) be the (mapped) baseband signal for the modulation scheme 64QAM. The mapping scheme for s 2 ( t ) is as shown in  FIG.  101   , and k is as represented by formula 85. Note that t is time. In the present embodiment, description is made taking the time domain as an example. 
     The power changer ( 9901 A) receives a (mapped) baseband signal  307 A for the modulation scheme 16QAM and the control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be v, the power changer outputs a signal ( 9902 A) obtained by multiplying the (mapped) baseband signal  307 A for the modulation scheme 16QAM by v. 
     The power changer ( 9901 B) receives a (mapped) baseband signal  307 B for the modulation scheme 64QAM and a control signal ( 9900 ) as input. Letting a value for power change set based on the control signal ( 9900 ) be u, the power changer outputs a signal ( 9902 B) obtained by multiplying the (mapped) baseband signal  307 B for the modulation scheme 64QAM by u. Then, let u=v×w (w&lt;1.0). 
     Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. ej θ(t) , formula 87 shown above is satisfied. 
     Therefore, a ratio of the average power for 64QAM to the average power for 16QAM is set to v 2 :u 2 =v 2 :v 2 ×w 2 =1:w 2 . With this structure, the reception device is in a reception condition as shown in  FIG.  98   . Therefore, data reception quality is improved in the reception device. 
     In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point. 
     The above describes that the values v and u for power change are set based on the control signal ( 9900 ). The following describes setting of the values v and u for power change based on the control signal ( 9900 ) in order to improve data reception quality in the reception device in detail. 
     Example 6-1 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a block length (the number of bits constituting one coded block, and is also referred to as the code length) for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of block lengths for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected block length for the error correction codes described above. The power changer ( 9901 B) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected block length indicated by the control signal ( 9900 ). Here, values for power change set according to the block length X are referred to as v LX  and u LX . 
     For example, when 1000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1000 . When 1500 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L1500 . When 3000 is selected as the block length, the power changer ( 9901 A) sets a value for power change to v L3000 . 
     On the other hand, when 1000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1000 . When 1500 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L1500 . When 3000 is selected as the block length, the power changer ( 9901 B) sets a value for power change to u L3000 . 
     In this case, for example, by setting v L1000 , v L1500  and v L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u L1000 , u L1500  and u L3000  so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u L1000 =u L1500  may be satisfied, and v L1000 =v L1500  may be satisfied. What is important is that two or more values exist in a set of v L1000 , v L1500  and v L3000 , and that two or more values exist in a set of u L1000 , u L1500  and u L1000 ). Note that, as described above, v LX  and u LX  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values u LX  for power change when the code length is set, and performs power change. Another important point is that two or more values v LX  for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values v LX  for power change when the code length is set, and performs power change. 
     Example 6-2 
     The following describes a scheme of setting the average power of s 1  and s 2  according to a coding rate for the error correction codes used to generate s 1  and s 2  when the transmission device supports a plurality of coding rates for the error correction codes. 
     Examples of the error correction codes include block codes such as Turbo codes or Duo-Binary Turbo codes using tail-biting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two systems. The encoded data having been distributed to the two systems is modulated in the modulation scheme for s 1  and in the modulation scheme for s 2  to generate the (mapped) baseband signals s 1 ( t ) and s 2 ( t ). 
     The control signal ( 9900 ) is a signal indicating the selected coding rate for the error correction codes described above. The power changer ( 9901 A) sets the value v for power change according to the control signal ( 9900 ). Similarly, the power changer ( 9901 B) sets the value u for power change according to the control signal ( 9900 ). 
     The present invention is characterized in that the power changers ( 9901 A and  9901 B) respectively set the values v and u for power change according to the selected coding rate indicated by the control signal ( 9900 ). Here, values for power change set according to the coding rate rx are referred to as v rx  and u rx . 
     For example, when r 1  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r1 . When r 2  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r2 . When r 3  is selected as the coding rate, the power changer ( 9901 A) sets a value for power change to v r3 . 
     Also, when r 1  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r1 . When r 2  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r2 . When r 3  is selected as the coding rate, the power changer ( 9901 B) sets a value for power change to u r3 . 
     In this case, for example, by setting v r1 , v r2  and v r3  so as to be different from one another, a high error correction capability can be achieved for each code length. Similarly, by setting u r1 , u r2  and u r3  so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, v r1 =v r2  may be satisfied, and u r1 =u r2  may be satisfied. What is important is that two or more values exist in a set of v r1 , v r2  and v r3 , and that two or more values exist in a set of u r1 , u r2  and u r3 ). Note that, as described above, v rx  and u rx  are set so as to satisfy the ratio of the average power 1:w 2 . 
     Also, note that, as examples of r 1 , r 2  and r 3  described above, coding rates ½, ⅔ and ¾ are considered when the error correction code is the LDPC code. 
     Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. One important point is that two or more values u rx  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values up, for power change when the coding rate is set, and performs power change. Another important point is that two or more values v rX  for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values v rX  for power change when the coding rate is set, and performs power change. 
     Example 6-3 
     In order for the reception device to achieve excellent data reception quality, it is important to implement the following. 
     The following describes a scheme of setting the average power (average values) of s 1  and s 2  according to a modulation scheme used to generate s 1  and s 2  when the transmission device supports a plurality of modulation schemes. 
     Here, as an example, a case where the modulation scheme for s 1  is fixed to 16QAM and the modulation scheme for s 2  is changed from 64QAM to QPSK by the control signal (or can be set to either 16QAM or QPSK) is considered. In a case where the modulation scheme for s 1  is 16QAM, the mapping scheme for s 1 ( t ) is as shown in  FIG.  95   , and g is represented by formula 79 in  FIG.  95   . In a case where the modulation scheme for s 2  is 64QAM, the mapping scheme for s 2 ( t ) is as shown in  FIG.  101   , and k is represented by formula 85 in  FIG.  101   . Also, in a case where the modulation scheme for s 2 ( t ) is QPSK, the mapping scheme for s 2 ( t ) is as shown in  FIG.  96   , and h is represented by formula 78 in  FIG.  96   . 
     In  FIG.  100   , when the modulation scheme for s 1  is 16QAM and the modulation scheme for s 2  is 64QAM, assume that v=α and u=α×w 64 . In this case, the ratio between the average power of 64QAM and the average power of 16QAM is v 2 :u 2 =α 2 :α 2 ×w 64   2 =1:w 64   2 . 
     In  FIG.  100   , when the modulation scheme for s 1  is 16QAM and the modulation scheme for s 2  is QPSK, assume that v=β and u=β×w 4 . In this case, the ratio between the average power of 64QAM and the average power of QPSK is v 2 :u 2 =β 2 :β 2 ×w 4   2 =1:w 4   2 . In this case, according to the minimum Euclidean distance relationship, the reception device achieves a high data reception quality when w 4 &lt;w 64 , regardless of whether the modulation scheme for s 2  is 64QAM or QPSK. 
     Note that although “the modulation scheme for s 1  is fixed to 16QAM” in the description above, it is possible that “the modulation scheme for s 2  is fixed to 16QAM and the modulation scheme for s 1  is changed from 64QAM to QPSK (set to either 16QAM or QPSK)”, w 4 &lt;w 64  should be fulfilled. (The same as described in Example 4-3). (Note that the value used for the multiplication for the power change in the case of 16QAM is u=α×w m , the value used for the multiplication for the power change in the case of QPSK is u=β×w 4 , the value used for the power change in the case of 64QAM is v=a when the selectable modulation scheme is 16QAM and v=β when the selectable modulation scheme is QPSK). Also, when the set of (the modulation scheme for s 1 , the modulation scheme for s 2 ) is selectable from the sets of (16QAM, 64QAM), (64QAM, 16QAM), (16QAM, QPSK) and (QPSK, 16QAM), w 4 &lt;w 64  should be fulfilled. 
     The following describes a case where the above-mentioned description is generalized. 
     For generalization, assume that the modulation scheme for s 1  is fixed to a modulation scheme C with which the number of signal points in the I-Q plane is c. Also assume that the modulation scheme for s 2  is selectable from a modulation scheme A with which the number of signal points in the I-Q plane is a and a modulation scheme B with which the number of signal points in the I-Q plane is b (c&gt;b&gt;a). In this case, when the modulation scheme for s 2  is set to the modulation scheme A, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme A, is 1:w a   2 . Also, when the modulation scheme for s 2  is set to the modulation scheme B, assume that ratio between the average power of the modulation scheme for s 1 , which is the modulation scheme C, and the average power of the modulation scheme for s 2 , which is the modulation scheme B, is 1:w b   2 . If this is the case, the reception device achieves a high data reception quality when w a &lt;w b  is fulfilled. 
     Note that although “the modulation scheme for s 1  is fixed to C” in the description above, even when “the modulation scheme for s 2  is fixed to the modulation scheme C and the modulation scheme for s 1  is changed from the modulation scheme A to the modulation scheme B (set to either the modulation scheme A or the modulation scheme B), the average powers should fulfill w a &lt;w b . (If this is the case, as with the description above, when the average power of the modulation scheme is C, the average power of the modulation scheme A is w a   2 , and the average power of the modulation scheme B is w b   2 .) Also, when the set of (the modulation scheme for s 1  and the modulation scheme for s 2 ) is selectable from the sets of (the modulation scheme C and the modulation scheme A), (the modulation scheme A and the modulation scheme C), (the modulation scheme C and the modulation scheme B) and (the modulation scheme B and the modulation scheme C), the average powers should fulfill w a &lt;w b . 
     In the present description including “Embodiment 1”, and so on, the power consumption by the transmission device can be reduced by setting α=1 in the formula 36 representing the precoding matrices used for the scheme for regularly changing the phase. This is because the average power of z 1  and the average power of z 2  are the same even when “the average power (average value) of s 1  and the average power (average value) of s 2  are set to be different when the modulation scheme for s 1  and the modulation scheme for s 2  are different”, and setting α=1 does not result in increasing the PAPR (Peak-to-Average Power Ratio) of the transmission power amplifier provided in the transmission device. 
     However, even when α≠1, there are some precoding matrices that can be used with the scheme that regularly changes the phase and have limited influence to PAPR. For example, when the precoding matrices represented by formula 36 in Embodiment 1 are used to achieve the scheme for regularly changing the phase, the precoding matrices have limited influence to PAPR even when α≠1. 
     Operations of the Reception Device 
     Subsequently, explanation is provided of the operations of the reception device. Explanation of the reception device has already been provided in Embodiment 1 and so on, and the structure of the reception device is illustrated in  FIGS.  7 ,  8 ,  9 ,  86 ,  87  and  88   , for instance 
     According to the relation illustrated in  FIG.  5   , when the transmission device transmits modulated signals as introduced in  FIGS.  99  and  100   , one relation among the two relations denoted by the two formulas below is satisfied. Note that in the two formulas below, r 1 ( t ) and r 2 ( t ) indicate reception signals, and h 11 ( t ), h 12 ( t ), h 21 ( t ), and h 22 ( t ) indicate channel fluctuation values. 
     In the case of Example 1, Example 2 and Example 3, the following relationship shown in formula 89 is derived from  FIG.  5   . 
     
       
         
           
             
               
                 
                   
                       
                   
                   ⁢ 
                   
                     [ 
                     
                       Math 
                       . 
                       
                           
                       
                       ⁢ 
                       89 
                     
                     ] 
                   
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
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                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     89 
                   
                   ) 
                 
               
             
           
         
       
     
     Also, as explained in Example 1, Example 2, and Example 3, the relationship may be as shown in formula 90 below: 
     
       
         
           
             
               
                 
                   
                       
                   
                   ⁢ 
                   
                     [ 
                     
                       Math 
                       . 
                       
                           
                       
                       ⁢ 
                       90 
                     
                     ] 
                   
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
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                   ( 
                   
                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     90 
                   
                   ) 
                 
               
             
           
         
       
     
     The reception device performs demodulation (detection) (i.e. estimates the bits transmitted by the transmission device) by using the relationships described above (in the same manner as described in Embodiment 1 and so on). 
     In the case of Example 4, Example 5 and Example 6, the following relationship shown in formula 91 is derived from  FIG.  5   . 
     
       
         
           
             
               
                 
                   
                       
                   
                   ⁢ 
                   
                     [ 
                     
                       Math 
                       . 
                       
                           
                       
                       ⁢ 
                       90 
                     
                     ] 
                   
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
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                     formula 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     91 
                   
                   ) 
                 
               
             
           
         
       
     
     Also, as explained in Example 3, Example 4, and Example 5, the relationship may be as shown in formula 92 below: 
     
       
         
           
             
               
                 
                   
                       
                   
                   ⁢ 
                   
                     [ 
                     
                       Math 
                       . 
                       
                           
                       
                       ⁢ 
                       92 
                     
                     ] 
                   
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
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                   ( 
                   
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                     ⁢ 
                     
                         
                     
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                     92 
                   
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     The reception device performs demodulation (detection) (i.e. estimates the bits transmitted by the transmission device) by using the relationships described above (in the same manner as described in Embodiment 1 and so on). 
     Note that although Examples 1 through 6 show the case where the power changer is added to the transmission device, the power change may be performed at the stage of mapping. 
     As described in Example 1, Example 2, and Example 3, and as particularly shown in formula 89, the mapper  306 B in  FIG.  3    and  FIG.  4    may output u×s 2 ( t ), and the power changer may be omitted in such cases. If this is the case, it can be said that the scheme for regularly changing the phase is applied to the signal s 1 ( t ) after the mapping and the signal u×s 2 ( t ) after the mapping, the modulated signal after precoding. 
     As described in Example 1, Example 2, and Example 3, and as particularly shown in formula 90, the mapper  306 A in  FIG.  3    and  FIG.  4    may output u×s 1 ( t ), and the power changer may be omitted in such cases. If this is the case, it can be said that the scheme for regularly changing the phase is applied to the signal s 2 ( t ) after the mapping and the signal u×s 1 ( t ) after the mapping, the modulated signal after precoding. 
     In Example 4, Example 5, and Example 6, as particularly shown in formula 91, the mapper  306 A in  FIG.  3    and  FIG.  4    may output v×s 1 ( t ), and the mapper  306 B may output u×s 2 ( t ), and the power changer may be omitted in such cases. If this is the case, it can be said that the scheme for regularly changing the phase is applied to the signal v×s 1 ( t ) after the mapping and the signal u×s 2 ( t ) after the mapping, the modulated signals after precoding. 
     In Example 4, Example 5, and Example 6, as particularly shown in formula 92, the mapper  306 A in  FIG.  3    and  FIG.  4    may output u×s 1 ( t ), and the mapper  306 B may output v×s 2 ( t ), and the power changer may be omitted in such cases. If this is the case, it can be said that the scheme for regularly changing the phase is applied to the signal u×s 1 ( t ) after the mapping and the signal v×s 2 ( t ) after the mapping, the modulated signals after precoding. 
     Note that F shown in formulas 89 through 92 denotes precoding matrices used at time t, and y(t) denotes phase changing values. The reception device performs demodulation (detection) by using the relationships between r 1 ( t ), r 2 ( t ) and s 1 ( t ), s 2 ( t ) described above (in the same manner as described in Embodiment 1 and so on). However, distortion components, such as noise components, frequency offset, channel estimation error, and the likes are not considered in the formulas described above. Hence, demodulation (detection) is performed with them. Regarding the values u and v that the transmission device uses for performing the power change, the transmission device transmits information about these values, or transmits information of the transmission mode (such as the transmission scheme, the modulation scheme and the error correction scheme) to be used. The reception device detects the values used by the transmission device by acquiring the information, obtains the relationships described above, and performs the demodulation (detection). 
     In the present embodiment, the switching between the phase changing values is performed on the modulated signal after precoding in the time domain. However, when a multi-carrier transmission scheme such as an OFDM scheme is used, the present invention is applicable to the case where the switching between the phase changing values is performed on the modulated signal after precoding in the frequency domain, as described in other embodiments. If this is the case, t used in the present embodiment is to be replaced with f (frequency ((sub) carrier)). 
     Accordingly, in the case of performing the switching between the phase changing values on the modulated signal after precoding in the time domain, z 1 ( t ) and z 2 ( t ) at the same time point is transmitted from different antennas by using the same (common/shared) frequency. On the other hand, in the case of performing the switching between the phase changing values on the modulated signal after precoding in the frequency domain, z 1 ( f ) and z 2 ( f ) at the same (common/shared) frequency is transmitted from different antennas at the same time point. 
     Also, even in the case of performing switching between the phase changing values on the modulated signal after precoding in the time and frequency domains, the present invention is applicable as described in other embodiments. The scheme pertaining to the present embodiment, which switches between the phase changing values on the modulated signal after precoding, is not limited the scheme which switches between the phase changing values on the modulated signal after precoding as described in the present Description. 
     Also, assume that processed baseband signals z 1 ( i ), z 2 ( i ) (where i represents the order in terms of time or frequency (carrier)) are generated by regular phase change and precoding (it does not matter which is performed first) on baseband signals s 1 ( i ) and s 2 ( i ) for two streams. Let the in-phase component I and the quadrature component Q of the processed baseband signal z 1 ( i ) be I 1 ( i ) and Q 1 ( i ) respectively, and let the in-phase component I and the quadrature component Q of the processed baseband signal z 2 ( i ) be I 2 ( i ) and Q 2 ( i ) respectively. In this case, the baseband components may be switched, and modulated signals corresponding to the switched baseband signal r 1 ( i ) and the switched baseband signal r 2 ( i ) may be transmitted from different antennas at the same time and over the same (common/shared) frequency by transmitting a modulated signal corresponding to the switched baseband signal r 1 ( i ) from transmit antenna  1  and a modulated signal corresponding to the switched baseband signal r 2 ( i ) from transmit antenna  2  at the same time and over the same (common/shared) frequency. Baseband components may be switched as follows.
         Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 ( i ) and Q 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 ( i ) and I 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 ( i ) and Q 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 ( i ) and Q 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 ( i ) and I 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 ( i ) and Q 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 ( i ) and I 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 ( i ) and I 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 ( i ) and I 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 ( i ) and Q 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 ( i ) and Q 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 ( i ) and I 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 ( i ) and Q 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 ( i ) and Q 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 ( i ) and I 2 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 ( i ) and Q 1 ( i ) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 ( i ) and I 1 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 ( i ) and I 2 ( i ) respectively.       

     In the above description, signals in two streams are processed and in-phase components and quadrature components of the processed signals are switched, but the present invention is not limited in this way. Signals in more than two streams may be processed, and the in-phase components and quadrature components of the processed signals may be switched. 
     In addition, the signals may be switched in the following manner. For example,
         Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 ( i ) and Q 2 ( i ) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 ( i ) and Q 1 ( i ) respectively.       

     Such switching can be achieved by the structure shown in  FIG.  55   . 
     In the above-mentioned example, switching between baseband signals at the same time (at the same (common/shared) frequency ((sub)carrier)) has been described, but the present invention is not limited to the switching between baseband signals at the same time. As an example, the following description can be made.
         Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 (i+v) and Q 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 (i+v) and I 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 (i+v) and Q 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 (i+v) and Q 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 (i+v) and I 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 1 (i+v) and Q 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 (i+v) and I 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 1 (i+v) and I 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 (i+v) and I 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 (i+v) and Q 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 (i+v) and Q 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 (i+v) and I 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 (i+v) and Q 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 (i+v) and Q 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 (i+v) and I 2 (i+w) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 (i+w) and Q 1 (i+v) respectively.   Let the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be Q 2 (i+w) and I 1 (i+v) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be Q 1 (i+v) and I 2 (i+w) respectively.       

     In addition, the signals may be switched in the following manner. For example,
         Let the in-phase component and the quadrature component of the switched baseband signal r 1 ( i ) be I 2 (i+w) and Q 2 (i+w) respectively, and the in-phase component and the quadrature component of the switched baseband signal r 2 ( i ) be I 1 (i+v) and Q 1 (i+w) respectively.       

     This can also be achieved by the structure shown in  FIG.  55   . 
       FIG.  55    illustrates a baseband signal switcher  5502  explaining the above. As shown, of the two processed baseband signals z 1 ( i )  5501 _ 1  and z 2 ( i )  5501 _ 2 , processed baseband signal z 1 ( i )  5501 _ 1  has in-phase component I 1 ( i ) and quadrature component Q 1 ( i ), while processed baseband signal z 2 ( i )  5501 _ 2  has in-phase component I 2 ( i ) and quadrature component Q 2 ( i ). Then, after switching, switched baseband signal r 1 ( i )  5503 _ 1  has in-phase component I r1 ( i ) and quadrature component Q r1 ( i ), while switched baseband signal r 2 ( i )  5503 _ 2  has in-phase component I r2 ( i ) and quadrature component Q r2 ( i ). The in-phase component I r1 ( i ) and quadrature component Q r1 ( i ) of switched baseband signal r 1 ( i )  5503 _ 1  and the in-phase component Ir 2 ( i ) and quadrature component Q r2 ( i ) of switched baseband signal r 2 ( i )  5503 _ 2  may be expressed as any of the above. Although this example describes switching performed on baseband signals having a common time (common ((sub-)carrier) frequency) and having undergone two types of signal processing, the same may be applied to baseband signals having undergone two types of signal processing but having different time (different ((sub-)carrier) frequencies). 
     The switching may be performed while regularly changing switching methods. 
     For example, 
     
         
         
           
             At time  0 ,
 
for switched baseband signal r 1 ( 0 ), the in-phase component may be I 1 ( 0 ) while the quadrature component may be Q 1 ( 0 ), and for switched baseband signal r 2 ( 0 ), the in-phase component may be I 2 ( 0 ) while the quadrature component may be Q 2 ( 0 );
 
             At time  1 ,
 
for switched baseband signal r 1 ( 1 ), the in-phase component may be I 2 ( 1 ) while the quadrature component may be Q 2 ( 1 ), and for switched baseband signal r 2 ( 1 ), the in-phase component may be I 1 ( 1 ) while the quadrature component may be Q 1 ( 1 ), and so on. In other words,
 
             When time is 2k (k is an integer),
 
for switched baseband signal r 1 (2k), the in-phase component may be I 1 (2k) while the quadrature component may be Q 1 (2k), and for switched baseband signal r 2 (2k), the in-phase component may be I 2 (2k) while the quadrature component may be Q 2 (2k).
 
             When time is 2k+1 (k is an integer),
 
for switched baseband signal r 1 (2k+1), the in-phase component may be I 2 (2k+1) while the quadrature component may be Q 2 (2k+1), and for switched baseband signal r 2 (2k+1), the in-phase component may be I 1 (2k+1) while the quadrature component may be Q 1 (2k+1).
 
             When time is 2k (k is an integer),
 
for switched baseband signal r 1 (2k), the in-phase component may be I 2 (2k) while the quadrature component may be Q 2 (2k), and for switched baseband signal r 2 (2k), the in-phase component may be I 1 (2k) while the quadrature component may be Q 1 (2k).
 
             When time is 2k+1 (k is an integer),
 
for switched baseband signal r 1 (2k+1), the in-phase component may be I 1 (2k+1) while the quadrature component may be Q 1 (2k+1), and for switched baseband signal r 2 (2k+1), the in-phase component may be I 2 (2k+1) while the quadrature component may be Q 2 (2k+1).
 
           
         
       
    
     Similarly, the switching may be performed in the frequency domain. In other words,
         When frequency ((sub) carrier) is 2k (k is an integer),
 
for switched baseband signal r 1 (2k), the in-phase component may be I 1 (2k) while the quadrature component may be Q 1 (2k), and for switched baseband signal r 2 (2k), the in-phase component may be I 2 (2k) while the quadrature component may be Q 2 (2k).
   When frequency ((sub) carrier) is 2k+1 (k is an integer),
 
for switched baseband signal r 1 (2k+1), the in-phase component may be I 2 (2k+1) while the quadrature component may be Q 2 (2k+1), and for switched baseband signal r 2 (2k+1), the in-phase component may be I 1 (2k+1) while the quadrature component may be Q 1 (2k+1).
   When frequency ((sub) carrier) is 2k (k is an integer),
 
for switched baseband signal r 1 (2k), the in-phase component may be I 2 (2k) while the quadrature component may be Q 2 (2k), and for switched baseband signal r 2 (2k), the in-phase component may be I 1 (2k) while the quadrature component may be Q 1 (2k).
   When frequency ((sub) carrier) is 2k+1 (k is an integer),
 
for switched baseband signal r 1 (2k+1), the in-phase component may be I 1 (2k+1) while the quadrature component may be Q 1 (2k+1), and for switched baseband signal r 2 (2k+1), the in-phase component may be I 2 (2k+1) while the quadrature component may be Q 2 (2k+1).
 
(Regarding Cyclic Q Delay)
       

     The following describes the application of the Cyclic Q Delay mentioned throughout the present disclosure. Non-Patent Literature 10 describes the overall concept of Cyclic Q Delay. The following describes a specific example of a generation method for the s 1  and s 2  signals when Cyclic Q Delay is used. 
       FIG.  102    illustrates an example of a signal point arrangement in the I-Q plane when the modulation scheme is 16-QAM. As shown, when the input bits are b 0 , b 1 , b 2 , and b 3 , the bits take on either a value of 0000 or a value of 1111. For example, when the bits b 0 , b 1 , b 2 , and b 3  are to be expressed as 0000, then signal point  10201  of  FIG.  102    is selected, a value of the in-phase component based on signal point  10201  is taken as the in-phase component of the baseband signal, and a value of the quadrature component based on signal point  10201  is taken as the quadrature component of the baseband signal. When the bits b 0 , b 1 , b 2 , and b 3  are to be expressed as a different value, the in-phase component and the quadrature component of the baseband signal are generated similarly. 
       FIG.  103    illustrates a sample configuration of a signal generator for generating modulated signals s 1 ( t ) (where t is time) (alternatively, s 1 ( f ), where f is frequency) and s 2 ( t ) (alternatively, s 2 ( f )) from (binary) data when the cyclic Q delay is applied. 
     A mapper  10302  takes data  10301  and a control signal  10306  as input, and performs mapping in accordance with the modulation scheme of the control signal  10306 . For example, when 16-QAM is selected as the modulation scheme, mapping is performed as illustrated in  FIG.  102   . The mapper then outputs an in-phase component  10303 _A and a quadrature component  10303 _B for the mapped baseband signal. No limitation is intended to the modulation scheme being 16-QAM, and the operations are similar for other modulation schemes. 
     Here, the data at time  1  corresponding to the bits b 0 , b 1 , b 2 , and b 3  from  FIG.  102    are respectively indicated as b 01 , b 11 , b 21 , and b 31 . The mapper  10302  outputs the in-phase component I 1  and the quadrature component Q 1  for the baseband signal at time  1 , according to the data b 0 , b 1 , b 2 , and b 3  at time  1 . Similarly, another mapper  10302  outputs the in-phase component I 2  and the quadrature component Q 2  and so on for the baseband signal at time  2 . 
     A memory and signal switcher  10304  takes the in-phase component  10303 _A and the quadrature component  10303 _B of the baseband signal as input and, in accordance with a control signal  10306 , stores the in-phase component  10303 _A and the quadrature component  10303 _B of the baseband signal, switches the signals, and outputs modulated signal s 1 ( t ) ( 10305 _A) and modulated signal s 2 ( t ) ( 10305 _B). The generation method for the modulated signals s 1 ( t ) and s 2 ( t ) is described in detail below. 
     As described elsewhere in the disclosure, precoding and phase changing are performed on the modulated signal s 1 ( t ) and s 2 ( t ). Here, as described elsewhere, signal processing involving phase change, power change, signal switching, and so on may be applied at any step. Thus, modulated signals r 1 ( t ) and r 2 ( t ), respectively obtained by applying the precoding and phase change to the modulated signals s 1 ( t ) and s 2 ( t ), are transmitted using the same (common) frequency band at the same (common) time. 
     Although the above description is given with respect to the time domain, s 1 ( t ) and s 2 ( t ) may be thought of as s 1 ( f ) and s 2 ( f ) (where f is the (sub-)carrier frequency) when a multi-carrier transmission scheme such as OFDM is employed. In contrast to the modulated signals s 1 ( f ) and s 2 ( f ), modulated signals r 1 ( f ) and r 2 ( f ) obtained using a precoding scheme in which the precoding matrix is regularly changed are transmitted at the same (common) time (r 1 ( f ) and r 2 ( f ) being, of course) signals of the same (common/shared) frequency band). Also, as described above, s 1 ( t ) and s 2 ( t ) may be treated as s 1 ( t,f ) and s 2 ( t,f ). 
     The following describes the generation method for modulated signals s 1 ( t ) and s 2 ( t ).  FIG.  104    illustrates a first example of a generation method for s 1 ( t ) and s 2 ( t ) when a cyclic Q delay is used. 
     Portion (a) of  FIG.  104    indicates the in-phase component and the quadrature component of the baseband signal obtained by the mapper  10302  of  FIG.  103   . As shown in  FIG.  87 A  and as described with reference to the mapper  10302  of  FIG.  103   , the mapper  10302  outputs the in-phase component and the quadrature component of the baseband signal such that in-phase component I 1  and quadrature component Q 1  occur at time  1 , in-phase component I 2  and quadrature component Q 2  occur at time  2 , in-phase component I 3  and quadrature component Q 3  occur at time  3 , and so on. 
     Portion (b) of  FIG.  104    illustrates a sample set of in-phase components and quadrature components for the baseband signal when signal switching is performed by the memory and signal switcher  10304  of  FIG.  103   . As shown, pairs of quadrature components are switched at each of time  1  and time  2 , time  3  and time  4 , and time  5  and time  6  (i.e., time  2   i +1 and time  2   i +2, i being a non-zero positive integer) such that, for example, the components at time  1  and t 2  are switched. 
     Accordingly, given that signal switching is not performed on the in-phase component of the baseband signal, the order thereof is such that in-phase component I 1  occurs at time  1 , in-phase component I 2  occurs at time  2 , baseband signal I 3  occurs at time  3 , and so on. 
     Then, signal switching is performed within the pairs of quadrature components for the baseband signal. Thus, quadrature component Q 2  occurs at time  1 , quadrature component Q 1  occurs at time  2 , quadrature component Q 4  occurs at time  3 , quadrature component Q 3  occurs at time  4 , and so on. 
     Portion (c) of  FIG.  104    indicates a sample configuration for modulated signals s 1 ( t ) and s 2 ( t ) before precoding, when the scheme applied involves precoding and phase changing. For example, as shown in portion (c), the baseband signal generated in portion (b) is alternately assigned to s 1 ( t ) and to s 2 ( t ). Thus, the first slot of s 1 ( t ) takes (I 1 , Q 2 ) and the first slot of s 2 ( t ) takes (I 2 , Q 1 ). Likewise, the second slot of s 1 ( t ) takes (I 3 , Q 4 ) and the second slot of s 2 ( t ) takes (I 4 , Q 3 ). This continues similarly. 
     Although  FIG.  104    describes an example with reference to the time domain, the same applies to the frequency domain (exactly as described above). In such cases, the descriptions pertain to s 1 ( f ) and  2 ( f ). 
     Then, N-slot precoded and phase changed modulated signals r 1 ( t ) and r 2 ( t ) are obtained after applying the precoding and phase change to the N-slot modulated signals s 1 ( t ) and s 2 ( t ). This point is described elsewhere in the present disclosure. 
       FIG.  105    illustrates a configuration that differs from that of  FIG.  103    and is used to obtain the N-slot s 1 ( t ) and s 2 ( t ) from  FIG.  104   . The mapper  10502  takes data and a control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  104   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates modulated signal s 1 ( t )( 10503 _A) and modulated signal s 2 ( t )( 10503 _B) from the mapped signal. Modulated signal (s 1 ( t ) ( 10503 _A) is identical to modulated signal  10305 _A from  FIG.  103   , and modulated signal s 2 ( t ) ( 10503 _B) is identical to modulated signal  10305 _B from  FIG.  103   . This is as indicated in portion (c) of  FIG.  104   . Accordingly, the first slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 1 , Q 2 ), the first slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 2 , Q 1 ), the second slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 3 , Q 4 ), the second slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 4 , Q 3 ), and so on. 
     The generation method for the first slot (I 1 , Q 2 ) of modulated signal s 1 ( t ) ( 10503 _A) and the first slot (I 2 , Q 1 ) of modulated signal s 2 ( t ) ( 10503 _B) by the mapper  10502  from  FIG.  105    is described below, as a supplement. 
     The data  10501  indicated in  FIG.  105    is made up of time  1  data b 01 , b 11 , b 21 , b 31  and of time  2  data b 02 , b 12 , b 22 , b 32 . The mapper  10502  of  FIG.  105    generates I 1 , Q 1 , I 2 , and Q 2  as described above using the data b 01 , b 11 , b 21 , b 31  and b 02 , b 12 , b 22 , and b 32 . Thus, the mapper  10502  of  FIG.  105    is able to generate the modulated signals s 1 ( t ) and s 2 ( t ) from I 1 , Q 1 , I 2 , and Q 2 . 
       FIG.  106    illustrates a configuration that differs from those of  FIGS.  103  and  105    and is used to obtain the N-slot s 1 ( t ) and s 2 ( t ) from  FIG.  104   . The mapper  10601 _A takes data  10501  and a control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  104   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates a modulated signal s 1 ( t ) ( 10503 _A) from the mapped signal. Similarly, the mapper  10601 _B takes data  10501  and a control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  104   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates a modulated signal s 2 ( t ) ( 10503 _B) from the mapped signal. 
     The data  10501  input to the mapper  10601 _A and the data  10501  input to the mapper  10601 _B are, of course, identical data. Modulated signal s 1 ( t ) ( 10503 _A) is identical to modulated signal  10305 _A from  FIG.  103   , and modulated signal s 2 ( t ) ( 10503 _B) is identical to modulated signal  10305 _B from  FIG.  6   . This is as indicated in portion (c) of  FIG.  104   . 
     Accordingly, the first slot of modulated signal s 1 ( t ) ( 10503 _A) takes (IL Q 2 ), the first slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 2 , Q 1 ), the second slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 3 , Q 4 ), the second slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 4 , Q 3 ), and so on. 
     The generation method for the first slot (I 1 , Q 2 ) of modulated signal s 1 ( t ) ( 10503 _A) by the mapper  10601 _A from  FIG.  106    is described below, as a supplement. The data  10501  indicated in  FIG.  106    are made up of time  1  data b 01 , b 11 , b 21 , b 31  and of time  2  data b 02 , b 12 , b 22 , b 32 . The mapper  10601 _A of  FIG.  106    generates I 1  and Q 2  as described above using the data b 01 , b 11 , b 21 , b 31  and b 02 , b 12 , b 22 , and b 32 . The mapper  10601 _A of  FIG.  106    then generates modulated signal s 1 ( t ) from I 1  and Q 2 . 
     The generation method for the first slot (I 2 , Q 1 ) of modulated signal s 2 ( t ) ( 10503 _B) by the mapper  10601 _B from  FIG.  106    is described below. The data  10501  indicated in  FIG.  106    are made up of time  1  data b 01 , b 11 , b 21 , b 31  and of time  2  data b 02 , b 12 , b 22 , b 32 . The mapper  10601 _B of  FIG.  106    generates I 2  and Q 1  as described above using the data b 01 , b 11 , b 21 , b 31  and b 02 , b 12 , b 22 , and b 32 . Thus, the mapper  10601 _B of  FIG.  106    is able to generate modulated signal s 2 ( t ) from I 2  and Q 1 . 
     Next,  FIG.  107    illustrates a second example that differs from the generation method of s 1 ( t ) and s 2 ( t ) from  FIG.  104    is given for a case where the cyclic Q delay is used. In  FIG.  107   , reference signs corresponding to elements found in  FIG.  104    are identical (i.e., the in-phase component and quadrature component of the baseband signal). 
     Portion (a) of  FIG.  107    indicates the in-phase component and the quadrature component of the baseband signal obtained by the mapper  10302  of  FIG.  103   . Portion (a) of  FIG.  107    is identical to portion (a) of  FIG.  104   . Explanations thereof are thus omitted. 
     Portion (b) of  FIG.  107    illustrates the configuration of the in-phase component and the quadrature component of the baseband signals s 1 ( t ) and s 2 ( t ) prior to signal switching. As shown, the baseband signal is allocated to s 1 ( t ) at times  2   i +1, and allocated to s 2 ( t ) at times  2   i +2 (i being a non-zero positive integer). 
     Portion (c) of  FIG.  107    illustrates a sample set of in-phase components and quadrature components for the baseband signal when signal switching is performed by the memory and signal switcher  10304  of  FIG.  103   . The main point of portion (c) of  FIG.  107    (and point of difference from portion (c) of  FIG.  104   ) is that signal switching occurs within s 1 ( t ) as well as s 2 ( t ). 
     Accordingly, in contrast to portion (b) of  FIGS.  107   , Q 1  and Q 3  of s 1 ( t ) are switched in portion (c) of  FIG.  107   , as are Q 5  and Q 7 . Also, in contrast to portion (b) of  FIGS.  107   , Q 2  and Q 4  of s 2 ( t ) are switched in portion (c) of  FIG.  107   , as are Q 6  and Q 8 . 
     Thus, the first slot of s 1 ( t ) has an in-phase component I 1  and a quadrature component Q 3 , and the first slot of s 2 ( t ) has an in-phase component I 2  and a quadrature component Q 4 . Also, the second slot of s 1 ( t ) has an in-phase component I 3  and a quadrature component Q 1 , and the second slot of s 2 ( t ) has an in-phase component I 4  and a quadrature component Q 4 . The third and fourth slots are as indicated in portion (c) of  FIG.  107   , and subsequent slots are similar. 
     Then, N-slot precoded and phase changed modulated signals r 1 ( t ) and r 2 ( t ) are obtained after applying the precoding and phase change to the N-slot modulated signals s 1 ( t ) and s 2 ( t ). This point is described elsewhere in the present disclosure. 
       FIG.  108    illustrates a configuration that differs from that of  FIG.  103    and is used to obtain the N-slot s 1 ( t ) and s 2 ( t ) from  FIG.  107   . The mapper  10502  takes data  10501  and a control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  107   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates modulated signal s 1 ( t )( 10503 _A) and modulated signal s 2 ( t )( 10503 _B) from the mapped signal. Modulated signal s 1 ( t ) ( 10503 _A) is identical to modulated signal  10305 _A from  FIG.  103   , and modulated signal s 2 ( t ) ( 10503 _B) is identical to modulated signal  10305 _B from  FIG.  6   . This is as indicated in portion (c) of  FIG.  107   . Accordingly, the first slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 1 , Q 3 ), the first slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 2 , Q 4 ), the second slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 3 , Q 1 ), the second slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 4 , Q 2 ), and so on. 
     The generation method for the first slot (I 1 , Q 3 ) of modulated signal s 1 ( t ) ( 10503 _A), the first slot (I 2 , Q 4 ) of modulated signal s 2 ( t ) ( 10503 _B), the second slot (I 3 , Q 1 ) of modulated signal s 1 ( t ) ( 10503 _A), and the second slot (I 4 , Q 2 ) of modulated signal s 2 ( t ) ( 10503 _B) by the mapper  10502  from  FIG.  108    is described below, as a supplement. 
     The data  10501  indicated in  FIG.  108    are made up of time  1  data b 01 , b 11 , b 21 , b 31 , time  2  data b 02 , b 12 , b 22 , b 32 , time  3  data b 03 , b 13 , b 23 , b 33 , and time  4  data b 04 , b 14 , b 24 , b 34 . The mapper  10502  of  FIG.  108    generates the aforementioned I 1 , Q 1 , I 2 , Q 2 , I 3 , Q 3 , I 4 , and Q 4  from the data b 01 , b 11 , b 21 , b 31 , b 02 , b 12 , b 22 , b 32 , b 03 , b 13 , b 23 , b 33 , b 04 . b 14 , b 24 , b 34 . Thus, the mapper  10502  of  FIG.  108    is able to generate the modulated signals s 1 ( t ) and s 2 ( t ) from I 1 , Q 1 , I 2 , Q 2 , I 3 , Q 3 , I 4 , and Q 4 . 
       FIG.  102    illustrates a configuration that differs from those of  FIGS.  103  and  108    and is used to obtain the N-slot s 1 ( t ) and s 2 ( t ) from  FIG.  107   . A distributor  10201  takes data  10501  and the control signal  10504  as input, distributes the data in accordance with the control signal  10504 , and outputs first data  10202 _A and second data  10202 _B. The mapper  10601 _A takes the first data  10202 _A and the control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  107   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates a modulated signal s 1 ( t )( 10503 _A) from the mapped signal. Similarly, the mapper  10601 _B takes second data  10202 _B and the control signal  10504  as input and, in accordance with the modulation scheme of the control signal  10504 , for example, performs mapping in consideration of the switching from  FIG.  107   , generates a mapped signal (i.e., in-phase components and quadrature components of the baseband signal) and generates a modulated signal s 2 ( t ) ( 10503 _B) from the mapped signal. 
     Accordingly, the first slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 1 , Q 3 ), the first slot of modulated signal s 2 ( t ) ( 10503 _B) takes (I 2 , Q 4 ), the second slot of modulated signal s 1 ( t ) ( 10503 _A) takes (I 3 , Q 1 ), the second slot of modulated signal s 2 ( t ) ( 10503 _B) takes ( 14 , Q 2 ), and so on. 
     The generation method for the first slot (I 1 , Q 3 ) of modulated signal s 1 ( t ) ( 10503 _A) and the first slot (I 3 , Q 1 ) of modulated signal s 2 ( t ) ( 10503 _B) by the mapper  10601 _A from  FIG.  109    is described below, as a supplement. The data  10501  indicated in  FIG.  109    are made up of time  1  data b 01 , b 11 , b 21 , b 31 , time  2  data b 02 , b 12 , b 22 , b 32 , time  3  data b 03 , b 13 , b 23 , b 33 , and time  4  data b 04 , b 14 , b 24 , b 34 . The distributor  10901  outputs the time  1  data b 01 , b 11 , b 21 , b 31  and the time  3  data b 03 , b 13 , b 23 , b 33 , as the first data  10902 _A, and outputs the time  2  data b 02 , b 12 , b 22 , b 32  and the time  4  data b 04 , b 14 , b 24 , b 34  as the second data  10902 _B The mapper  10601 _A of  FIG.  109    generates the first slot as (I 1 , Q 3 ) and the second slot as (I 3 , Q 1 ) from the data b 01 , b 11 , b 21 , b 31 , b 03 , b 13 , b 23 , b 33 . The third slot and subsequent slots are generated similarly. 
     The generation method for the first slot (I 2 , Q 4 ) of modulated signal s 2 ( t ) ( 10503 _B) and the second slot (I 4 , Q 2 ) by the mapper  10601 _B from  FIG.  109    is described below. The mapper  10601 _B from  FIG.  109    generates the first slot as (I 2 , Q 4 ) and the second slot as (I 4 , Q 2 ) from the time  2  data b 02 , b 12 , b 22 , b 32  and the time  4  data b 04 , b 14 , b 24 , b 34 . The third slot and subsequent slots are generated similarly. 
     Although two methods using cyclic Q delay are described above, when the signals are switched among slot pairs as per  FIG.  104   , the demodulator (detector) of the reception device is able to constrain the quantity of candidate signal points. This has the merit of reducing the scope of calculation (circuit scope). Also, when the signals are switched within s 1 ( t ) and s 2 ( t ), as per  FIG.  107   , the demodulator (detector) of the reception device encounters a large quantity of candidate signal points. However, time diversity gain (or frequency diversity gain when switching is performed with respect to the frequency domain) is available, which as the merit of enabling further improvements to the data reception quality. 
     Although the above description uses examples of a 16-QAM modulation scheme, no limitation is intended. The same applies to other modulation schemes, such as QPSK, 8-QAM, 32-QAM, 64-QAM, 128-QAM, 256-QAM and so on. 
     Also, the cyclic Q delay method is not limited to the two schemes given above. For example, either of the two schemes given above may involve switching either of the quadrature component or the in-phase component of the baseband signal. Also, while the above describes switching performed at two times (e.g., switching the quadrature components of the baseband signal at times  1  and  2 ), the in-phase components and (or) the quadrature components of the baseband signal may also be switched at a plurality of times. Accordingly, when the in-phase components and quadrature components of the baseband signal are generated and cyclic Q delay is performed as in  FIG.  104   , then the in-phase component of the baseband signal after cyclic Q delay at time i is Ii, and the quadrature component of the baseband signal after cyclic Q delay at time i is Qj (where i≠j). Alternatively, the in-phase component of the baseband signal after cyclic Q delay at time i is Ij, and the quadrature component of the baseband signal after cyclic Q delay at time i is Qi (where i≠j). Alternatively, the in-phase component of the baseband signal after cyclic Q delay at time i is Ij, and the quadrature component of the baseband signal after cyclic Q delay at time i is Qk (where i≠j, i≠k, j≠k). 
     The precoding and phase change are then applied to the modulated signals s 1 ( t ) (or s 1 ( f ), or s 1 ( t,f )) and s 2 ( t ) (or s 2 ( f ) or s 2 ( t,f )) obtained by applying the above-described cyclic Q delay. (Here, as described elsewhere, signal processing involving phase change, power change, signal switching, and so on may be applied at any step.) Here, the precoding and phase changing application method used on the modulated signal obtained with the cyclic Q delay may be any of the precoding and phase changing methods described in the present disclosure. 
     INDUSTRIAL APPLICABILITY 
     The present invention is widely applicable to wireless systems that transmit different modulated signals from a plurality of antennas, such as an OFDM-MIMO system. Also, the present invention is also applicable in a wired system having multiple connections (e.g., a power line communication system, a fibre-optic system, a digital subscriber line system, and so on) when MIMO transmission is used, and the modulated signals described in the present document are applied. The modulated signals may also be transmitted from a plurality of transmission locations. 
     REFERENCE SIGNS LIST 
     
         
         
           
               302 A,  302 B Encoders 
               304 A,  304 B Interleavers 
               306 A,  306 B Mappers 
               314  Signal processing scheme information generator 
               308 A,  308 B Weighting compositors 
               310 A,  310 B Wireless units 
               312 A,  312 B Antennas 
               317 A,  317 B Phase changers 
               402  Encoder 
               404  Distributor 
               504  # 1 ,  504  # 2  Transmit antennas 
               505  # 1 ,  505  # 2  Receive antennas 
               600  Weighting unit 
               701 _X,  701 _Y Antennas 
               703 _X,  703 _Y Wireless units 
               705 _ 1  Channel fluctuation estimator 
               705 _ 2  Channel fluctuation estimator 
               707 _ 1  Channel fluctuation estimator 
               707 _ 2  Channel fluctuation estimator 
               709  Control information decoder 
               711  Signal processor 
               803  Inner MIMO detector 
               805 A,  805 B Log-likelihood calculators 
               807 A,  807 B Deinterleavers 
               809 A, 809 B Log-likelihood ratio calculator 
               811 A,  811 B Soft-in/soft-out decoders 
               813 A,  813 B Interleavers 
               815  Memory 
               819  Coefficient generator 
               901  Soft-in/soft-out decoder 
               903  Distributor 
               1201 A,  1201 B OFDM-related processors 
               1302 A,  1302 A Serial-to-parallel converters 
               1304 A,  1304 B Reorderers 
               1306 A,  1306 B Inverse Fast Fourier Transform units 
               1308 A,  1308 B Wireless units