Patent Publication Number: US-9432228-B2

Title: Digital pre-distortion filter system and method

Description:
RELATED APPLICATION INFORMATION 
     The present application is a continuation of U.S. patent application Ser. No. 13/618,607, filed Sep. 14, 2012, now issued as U.S. Pat. No, 9,210,009, which claims priority under 35 U.S.C. 119(e) to U.S. Provisional Patent Application Ser. No. 61/535,208 filed Sep. 15, 2011, the disclosure of which is incorporated herein by reference in its entirety. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates in general to predistortion linearized communication systems and related methods. More particularly, the invention is directed to transmitters equipped with narrow bandwidth band-pass filters for terrestrial communication networks. 
     2. Description of the Prior Art and Related Background Information 
     Wireless network operators are facing ever shrinking spectrum challenges. In high density markets such as that found in larger urban environments, every available kHz of available spectrum has been allocated and used to provide voice and data traffic between mobile user equipment (“UE”) and the Base Station (“BS”). To increase wireless capacity, network operators are forced to add carriers and/or split cell sites but eventually are reaching interference limits or are unable to find a suitable location where another BS and its associated equipment (e.g., tower, equipment, antennas, etc) can be installed. Further complicating and limiting the amount of useful spectrum are regulatory interference requirements placed on wireless network operators. Regulatory requirements, such as those mandated by the FCC, stipulate the amount of harmful interference that can be tolerated from BS transmitters operating within its assigned frequency allocation to nearby communication services. In most cases, detected levels of harmful interference levels caused by BS transmitters in nearby communication spectrum must be substantially attenuated and not exceed prescribed levels. To keep the amount of harmful interference from spilling into adjacent communication bands, BS transmitters usually employ highly linear transmitters which keep distortion products at a minimum. Highly linear transmitters utilize power amplifiers which must maintain as linear operation as possible, and the power amplifier is designed to operate within its linear region given the range of possible input signal amplitudes. However if the input signal has an amplitude which causes the power amplifier to operate outside the linear region, the power amplifier introduces nonlinear components or distortion to the signal. Generally, power amplifiers are characterized as having a compression threshold, and input signals having amplitudes above compression threshold are clipped at the amplifier output. In addition to distorting the amplified input signal, the clipping or nonlinear amplification of the input signal generates spectral regrowth which can interfere with communication services in adjacent frequency bands. The problem of non-linear distortion is very common in wireless communications systems that provide high power amplification of transmit signals with very large peak to average power ratios (“PAR”). In one example of large PAR signals of a code division multiple access (“CDMA”) system, a single 1.25 MHz wide carrier can typically have a PAR of 11.3 dB. In another example orthogonal frequency division multiplexing (“OFDM”), multicarrier signals can have a PAR of up to 20 dB. 
     Unfortunately, the efficiency of the BS amplifier is inversely related to its linearity. To achieve a higher degree of linearity, the amplifiers are biased to operate in the class AB. Numerous techniques and amplifier topologies are used to maximize amplifier RF to DC efficiency, but linearity requirements mandated by modern wireless communication systems dictate the use of class AB modes or combination of AB and C. Consequently, a significant portion of DC power is dissipated by the amplifiers as heat which must be removed. Typically, BS amplifiers use heat sinks and fans to remove heat from RF power devices which further add cost, size, and weight to the base station equipment. Thus, there has been a great deal of effort to reduce the amount of heat generated by BS power amplifiers in a quest to improve amplifier efficiency without degrading amplifier linearity. 
     When a power amplifier (“PA”) is operated with CDMA (or similar signals) at its input, the PA will amplify the desired signal as well as generate unwanted intermodulation (“IM”) products. These IM products increase rapidly as the PA output is driven to its saturation point. To achieve the desired linearity at the PA output (without predistortion), the PA must be operated at backoff output power level from its saturation point (PSAT 3dB ). Unfortunately PA operation at backoff power levels limits PA&#39;s maximum useful output power level so that the full range of output signal dynamic range is well within the linear region of the PA transfer curve. However, such operating point negatively impacts the PA&#39;s efficiency. Efficiencies of 10% or less for conventionally constructed Class AB PAs are not unusual when operated with input signals having 8 to 9 dB peak-to-average ratio PAR while only marginally meeting system linearity requirements. 
     In view of recent developments in Digital Pre-Distortion (“DPD”), DPD has become a linearization method of choice for Class A/AB PAs transmitting 60 W average power and below. A DPD linearization approach lends itself in solving several previously un-attainable performance limitations such as exhibiting high efficiency with good linearity. These improvements stem from DPD operating point that allows PA&#39;s to operate close or even slightly above its P SAT  during peak signal transition. DPD usually use techniques where a correction signal is created and amplified along with input signal through the PA&#39;s in order to reduce the overall distortion at the output of the PA. A DPD can be an optimized CDMA signal (such as IS-95) which tends to have with a large PAR 9.7 dB (0.01% probability on the CCDF) for a single carrier CDMA with pilot, paging, sync and 6 traffic channels (Walsh codes 8-13). A single channel IS-95 has channel bandwidth 1.23 MHz and DPD generally is optimized to reduce third-order IM products. In such application, DPD can predistort PA so that resultant Adjacent Channel Leakage Ratio (“ACLR”) of 48 to 50 dBc at 885-kHz offset, which is typically 14 dB or better over PA ACLR performance operating without predistortion. Similarly, a DPD can be also be optimized to provide cancellation performance for a four carrier WCDMA input signal over a 20 MHz signal band. The DPD performance is usually hampered by the memory effects of the PA which potentially limit DPD effectiveness. The memory effects in a PA amplifier stage are defined as a change of the amplitude and phase in distortion components due to the previous signals. However, if the PA is designed with bias circuits that tend to reduce or limit memory effects, DPD can typically provide linearization for a four-carrier WCDMA signal that results in ACLR of 46 to 48 dBc range with 13 dB or better cancellation at 5-MHz offset. In most applications, PA efficiencies above 25 percent for Class AB PAs and to 40 percent or more for Doherty type PAs is achievable. 
     Given that DPD linearization can provide ACLR performance additional attenuation of IM products into adjacent spectrum can be achieved by providing a bandpass filter or in case of FDD system duplexer can be implemented in the output of the RRH. 
     In a typical FDD implementation, a cavity duplexer provides isolation between receiver and transmitter of the RRH as well as adequate degree of TX IM rejection into adjacent spectrum. Since a duplexer is generally required for FDD RRH, its inclusion introduces a host of issues, such as additional output insertion loss and filter pass band response ripple on the transmitted signal quality. The latter manifests itself in an operational situation where CDMA carrier is positioned close to filter roll-off or a band edge. This usually takes place when a network operator configures a CDMA (or WCDMA) carrier in its allocated frequency spectrum in close proximity to the band edge. When such carrier frequency is selected, the net effect of the filter roll-off characteristic impacts CDMA carrier flatness which in turn degrades Rho (“ρ”). The Rho of the BS is figure of merit which is a measure of modulation quality of the transmitted CDMA signal. A Rho of 1.0 is associated with an ideal transmitted signal, essentially all of the power in the CDMA carrier is being transmitted correctly without any multipath. The CDMA standard calls for Rho&gt;0.912 and, in practice, BS transmitter measurements Rho&gt;0.94 indicate a normal BS transmitter operation. 
     For BS operating W-CDMA air standard (per 3GPP) BS transmitter quality can be summarized in the table below: 
     
       
         
           
               
            
               
                   
               
               
                 3GPP Requirements 
               
            
           
           
               
               
               
            
               
                   
                 3GPP  
                 Requirement Limits 
               
               
                   
                   
               
               
                   
                 EVM 
                 17.5% 
               
            
           
           
               
               
               
               
            
               
                   
                 PCDE  
                 −33  
                 dB 
               
               
                   
                 ACLR1 
                 45  
                 dB 
               
               
                   
                 ACLR2 
                 50  
                 dB 
               
               
                   
                   
               
            
           
         
       
     
     The Error Vector Magnitude is a measure of the difference between the reference waveform and the measured waveform. This difference is called the error vector. Both waveforms pass through a matched Root Raised Cosine filter with bandwidth 3.84 MHz and roll-off α=0.22. Both waveforms are then further modified by selecting the frequency, absolute phase, absolute amplitude and chip clock timing so as to minimize the error vector. The EVM result is defined as the square root of the ratio of the mean error vector power to the mean reference power expressed as a percentage (“%”). The EVM measurement in the test set compares the received signal&#39;s IQ modulation characteristics to an ideal signal, as defined in 3GPP TS 34.121 section 5.13 and annex B. 
     Introduction of a narrow bandpass filter in the output of the BS PA results in CDMA carrier flatness degradation especially if the carrier is positioned close to filter roll off. Typical narrow band pass filters also tend to exhibit rapid group delay change near filter roll-off. The combination of amplitude and group delay variation introduced by the narrow band pass filter degrades Rho and EVM parameters. 
     Accordingly, a need exists to improve performance in communication systems. 
     SUMMARY OF THE INVENTION 
     In a first aspect, the present invention provides a predistortion linearized communication system, comprising an input receiving a digital communication signal, a predistorter coupled to the input, the predistorter outputting a predistorted signal, and an up-conversion module coupled to the predistorter receiving the predistorted signal and outputting an up-converted digital signal. The predistortion linearized communication system further comprises a digital to analog convertor coupled to the up-conversion module receiving the up-converted signal and outputting a RF signal, and a filter coupled to the digital to analog convertor receiving the RF signal and outputting a filtered RF signal. The predistorter outputs the predistorted signal based on the characteristics of the filter. 
     In a preferred embodiment, the predistorter comprises a finite impulse response filter. In an embodiment, the filter is a passband filter, the communication signal has one or more frequency bands and the predistorter provides a predistortion signal varying with frequency band location relative to the passband filter roll off characteristics. The predistortion linearized communication system preferably further comprises a filter group delay predistorter coupled between the up-conversion module and the digital to analog convertor, wherein the filter group delay predistorter provides an inverse group delay in the up-converted digital signal. The filter group delay predistorter preferably comprises an infinite impulse response filter. The infinite impulse response filter is preferably an all-pass infinite impulse response filter. The filter may include a combiner cavity filter receiving the RF signal and providing an output RF signal and the predistorter compensates for cavity filter insertion loss characteristics. The predistortion linearized communication system preferably further comprises a channel filtering and pulse shaping module receiving the input signal and providing baseband filtering to the input signal. The predistortion linearized communication system preferably further comprises an interpolation circuit coupled between the filter passband predistorter and the up-conversion module. 
     In another aspect, the present invention provides a predistortion linearized communication system for amplifying a digital communication signal, comprising an input receiving a multi-carrier digital communication signal, an input processing module coupled to the input and receiving the multi-carrier digital communication signal, the input processing module providing a plurality of sampled input signals. A plurality of digital up-converter circuit paths each receive a corresponding sampled input signal from the input processing module, each digital up-converter circuit path comprising a filter passband predistorter, the filter passband predistorter receiving a corresponding sampled input signal and outputting a predistorted signal, and an up-conversion module coupled to the filter passband predistorter receiving the predistorted signal and outputting up-converted digital signal. The predistortion linearized communication system further comprises a summer module receiving the up-converted digital signals from each of the digital up-convertor circuit paths and providing a multi-carrier composite signal, a filter group delay predistorter receiving the multi-carrier composite signal and providing a group delay compensated signal, and a digital to analog convertor coupled to the filter group delay predistorter receiving the group delay compensated signal and outputting a RF signal. A passband filter is coupled to the digital to analog convertor receiving the RF signal and outputting a passband RF signal. Each of the filter passband predistorters outputs the predistorted signals based on the characteristics of the passband filter relative to the multi-carrier digital communication signal. 
     In a preferred embodiment, the predistortion linearized communication system further comprises a combiner cavity filter receiving the passband RF signal and providing an output RF signal. Each of the digital up-convertor circuit paths preferably further comprises a channel filtering and pulse shaping module receiving the corresponding sampled input signal and providing baseband filtering to the sampled input signal. Each of the digital up-convertor circuit paths preferably further comprises an interpolation circuit coupled between the filter passband predistorter and the up-conversion module. Each of the filter passband predistorters preferably comprises a finite impulse response filter. Each of the finite impulse response filters preferably has 5 taps. The filter group delay predistorter preferably comprises an infinite impulse response filter. Each of the filter group delay predistorters preferably comprise an all-pass infinite impulse response filter. The predistortion linearized communication system preferably further comprises an uplink signal path coupled to the combiner cavity filter receiving uplink signals. 
     In another aspect, the present invention provides a method for predistortion linearization of a transmitter. The method comprises receiving a digital communication signal, providing a predistorted signal based on the digital communication signal employing digital predistortion coefficients, and providing an up-converted digital signal based on the predistorted signal. The method further comprises providing an RF signal based on the up-converted signal, and providing passband filtering by a passband filter on the RF signal. The digital predistortion coefficients are based on the characteristics of the passband filter and the communication signal frequency. 
     In a preferred embodiment the method for predistortion linearization further comprises providing group delay predistortion compensation on the up-converted signal. 
     Further features and aspects of the invention are set out in the following detailed description. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block schematic diagram of a digitally pre-distortion controlled transceiver. 
         FIG. 2  is a block schematic diagram of exemplary functional elements used in the pre-distortion system. 
         FIG. 2A  is a block schematic diagram of a Passband predistorter implemented with a complex FIR (“finite impulse response”) filter structure in an embodiment. 
         FIG. 2B  is a block schematic diagram of a Filter Group Delay Predistorter module implemented with IIR (“infinite impulse response”) all-pass filter structure in an embodiment. 
         FIG. 3  is a representation of a frequency response plot of output band pass filter and compensation provided by DPD. 
         FIG. 4  is a representation of calculated amplitude DPD required for low frequency positioned CDMA carrier depending on the CF 1  offset from the low or high band edge. 
         FIG. 5  is a representation of simulated amplitude and group delay characteristics (S 21 ) of a narrow band-pass filter (Cellular  850 ). 
         FIG. 6  is a representation of simulated amplitude and group delay characteristic low frequency detail of a narrow band-pass filter (Cellular  850 ). 
         FIG. 7  is a simulation of the FPPD effect on the output spectrum flatness. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     One or more embodiments solve the above described negative effects of the narrow band pass filters in the output of a BS PA and improves the output spectrum. In a preferred embodiment, this is achieved by digitally predistorting baseband signals in the Digital Up-Conversion (“DUC”) chain of the predistorter depending on the CDMA (“WCDMA”) carrier position (“Fc”) relative to filter roll-off characteristic. 
     One or more embodiments are described below with reference to the figures. With reference to  FIG. 1 , a DPD linearized transceiver  10 , which is also known as a remote radio head (“RRH”), for terrestrial wireless communication system is shown by way of a general system block diagram. As such it can support a number of air interface formats such as UMTS, CDMA-2000, IS-95, GSM, GPRS, WiMAX, HSPA, LTE, and other formats. Generally, an RRH can be deployed in near proximity of the antenna  308  to minimize transmission line losses between output combiner  306  and antenna  308 . The RRH is provided with power source connection and data connection to a base station server (“BSS,” not shown) which is connected to a core network. In common practice, a 2G/3G/4G base station is connected to the RRHs with a fiber optical connection. Either CPRI or OBSAI (other formats can be used) may be used to carry RF and operational control data to and from the RRH to provide wireless coverage in a dedicated geographic area. The RRH is equipped with a suitable I/O Unit  50  to provide flexible digital interfacing and processing functions between BSS and RRH. Microcontroller unit (“MCU”)  60  is provided to perform data routing, control, and configuration duties within the RRH. Numerous architectures and implementations can be successfully used for MCU  60  suitable for RRH operation. In general, the MCU  60  is provided with non volatile memory storage such as ROM  70  where operating system, programs and fixed parameters maybe stored along with a SDRAM  80  used for real time computational requirements of MCU  60 . For DPD functions along with uplink (“UL”) and observation path receiver decoding are provided in a field programmable gate array (“FPGA”)  90 . It should be noted that FPGA  90  can be replaced with a custom designed application specific integrated circuit (“ASIC”) which may incorporate many if not all functions programmed into FPGA as a part of its circuits. Unfortunately, a fully custom ASIC precludes any subsequent changes to algorithms or control flow changes offered by conventional FPGA. Insomuch, recent development of new class of ASIC have made transition from fully programmable FPGA to semi custom (or fully custom) ASIC easier by providing a combination ASIC-FPGA IC allowing DPD implementation some degree of freedom in case if changes are required. Detailed description of the FPGA  90  will be covered later on. 
     Now with reference to  FIGS. 1 and 2 , the FPGA  90  provides down link (“DL”) outputs  142  to TX module  302  of the digital to RF transceiver  300 . It should be noted that RF transceiver  300  can be equipped to provide multiple transmit path, such as to support 2×2 or 4×4 MIMO, but only one transmit path is shown for the sake of clarity. Furthermore, other well known features of DPD such as an observation receive path which is used by DPD to correct the output nonlinearity in the PA output signal have been omitted. The output of the TX chain is coupled into output duplexer  306  such that RRH TX signals are further coupled to a common transmit receive antenna  308 . Coincidently, received uplink (“UL”) path signals are coupled from antenna  308  through the duplexer  306  to RX chain  310 . Again, as in case of TX chain, only one RX chain is shown, but it shall be understood that a plurality of RX chains can be used to support 2×2, 4×4 MIMO or auxiliary sector augmentation receivers can be implemented as required. The digital output  144  of RX chain  310  is coupled to RX FPGA  146  portion of the FPGA  90  such that received UL signals are processed and routed via RRH BSS interface to a core network. 
     The digital functionality of the RRH has been implemented in the FPGA  90  which contains logic elements (“LEs”) multipliers and embedded memory. Processed RF baseband data is received from I/O unit  50  into input of one of several digital upconverter chains (“DUC”)  100 . Depending on RRH deployment configuration, there maybe one or more DUC, each used with a specific RF carrier frequency. The received composite RF baseband  102  data is coupled into I/Q input processing module  104  which selectively routes I/Q samples to respective DUC  100 - 1  to  4  depending on RF carrier frequency and provides channel filtering designed for bandwidth of 1.25 MHz. Alternative channel bandwidth filtering is readily possible depending on transmission signal standard. Outputs  106 - 1  to  4  of CPRI I/Q input processing module  104  are coupled into channel filter—pulse shaper module  108 . The Channel Filtering+pulse shaping module  108  provides baseband filter characteristics specified in the CDMA 2000 standard. The output  110  of the Channel Filter—pulse shaper module  108  is coupled to Filter Passband Predistorter (“FPPD”) module  112  which provides inverse amplitude waveform predistortion based on carrier location relative to filter roll-off characteristic. This will be described later on. It should be noted that FPPD module  112  can be positioned after interpolation  116 , but this will result in increased number of logic gates needed to implement FPPD  112 . The output  114  of FFPD module  112  is coupled into interpolation module  116  with its output  118  coupled to numerically controlled oscillator (“NCO”) up conversion module  120 . The output of each DUC  100 - 1  to n  126 - 1  to  126 - 4  is routed  128 - 1  to n to inputs  130 - 1  to  130 - 4  of a summer module  132  where a multi-carrier composite signal  134  is further processed through Filter Group Delay Predistorter  136  module (“FGDPD”). Filter Group Delay Predistorter  136  module provides inverse group delay characteristic into a multicarrier signal stream so as to remove group delay deviation present in the output combiner duplexer  306 . Filter Group Delay Predistorter  136  module output  138  is coupled into crest factor (“CFR”) and Digital predistortion (“DPD”) module  140  before being coupled  142  into Digital to Analog Converter of the TX chain  302  to be converted and amplified to a desired RF signal. 
     With reference to  FIG. 3  a two carrier IS-95 CDMA deployment scenario is presented by way of example, however deployment of two adjacent CDMA carriers is not a limiting factor and any number or types (i.e. IS-95 with LTE) of carriers can be deployed. In  FIG. 5 , a TX to ANT duplexer filter characteristics (S 21  loss and S 21  group delay) are depicted for operation in DL TX band allocated between 862.275-868.725 MHz, while UL RX signals are 817 to 824 MHz range. Rejection of TX signals in 824-861.5 MHz is specified at 50 dB Min. 
     
       
         
           
               
               
               
               
             
               
                   
                   
               
             
            
               
                   
                 TX to ANT 
                 817-824 
                 100 dB Min 
               
               
                   
                 TX to ANT 
                 824-861.5 
                  50 dB Min 
               
               
                   
                 TX to ANT 
                 875-880 
                  20 dB Min 
               
               
                   
                 TX to ANT 
                 880-900 
                  35 dB Min 
               
               
                   
                 TX to ANT 
                 900-2700 
                  65 dB Min 
               
               
                   
                   
               
            
           
         
       
     
     
       
         
           
               
               
               
               
               
             
               
                   
                   
               
             
            
               
                   
                 Tx Ripple 
                 TX to ANT  
                 862.275-863.525 
                  2.0 dB Max 
               
               
                   
                 Tx Ripple 
                 TX to ANT 
                 863.525-868.725 
                 0.45 dB Max 
               
               
                   
                   
               
            
           
         
       
     
     When such duplexer  306  is introduced in the output of the TX chain  302 , the Rho (“ρ”) of the CDMA transmitted signal, especially at 862.90 MHz, is substantially impaired. This impairment is caused by filter S 21  insertion loss and group delay ripple between 862.275 and 863.525 MHz. To compensate for the filter characteristics, embodiments utilize digital predistortion means to compensate for filter&#39;s S 21  insertion loss and group delay ripple. Each DUC  100 - 1  to n includes digital filter passband predistorter module  112  in it signal processing chain. Since the carrier frequency processed by a specific DUC  100 - 1  to n is known the amount of amplitude correction provided by filter passband predistorter module  112  can be calculated based on filter roll-off characteristic and frequency of the CDMA carrier. An array of amplitude correction curves are depicted in  FIG. 4  which can be selected by FPPD  112  depending on operational requirements. 
     As shown in  FIG. 2A , the FPPD  112  has a complex finite impulse response filter structure and takes complex weighted input samples x(n)  110  and has complex weighted output samples y(n)  114  with a transfer function H(z) such that y(n) is equal to x(n) convolved with H(z) following the equation below. Where:
         n is the number of taps (5 in this example),   b are the complex weighted coefficients, and   z is a delay by k samples (k is integer).       

     
       
         
           
             
               H 
               ⁡ 
               
                 ( 
                 z 
                 ) 
               
             
             = 
             
               
                 ∑ 
                 
                   k 
                   = 
                   0 
                 
                 n 
               
               ⁢ 
               
                   
               
               ⁢ 
               
                 
                   b 
                   ⁡ 
                   
                     ( 
                     k 
                     ) 
                   
                 
                 ⁢ 
                 
                   z 
                   
                     - 
                     k 
                   
                 
               
             
           
         
       
     
       FIG. 2A  depicts an exemplary Passband predistorter  112 . The input x(n)  110  is coupled to the input of the Z −1  unit  202   a  and the multiplier  204   a . Multiplier  204   a  multiplies the input  110  and coefficient b( 0 )  208   a  and outputs the product to adder  206 . The output of Z −1  unit  202   a  is coupled to the input of Z −1  unit  202   b  and multiplier  204   b . Multiplier  204   b  multiplies the input of Z −1  unit  202   b  and coefficient b( 1 )  208   b , and outputs the product to adder  206 . The circuit may be expanded to include additional taps. Z −1  unit  202   n  outputs the signal to multiplier  204 ( n+ 1) which multiplies the signal with b(n)  208   n  and provides an output to adder  206 . Adder  206  sums the respective signals and provides the output y(n)  114 . 
     It shall be understood that numerous methods known to the skilled in the art are available to generate such filter compensation curves, together with adjustments for duplexer to duplexer variations, temperature compensation and the like, as required, to maintain acceptable Rho performance. 
     One or more embodiments preferably utilizes 5 tap FIR predistortion correction filter in the DUC  100  chain to normalize amplitude response introduced by the narrow band-pass filter  304  and to a lesser degree output duplexer  306 . Examples will be made with reference to a DPD linearized PA operating in cellular 850 band specifically between 862.275 to 863.525 MHz, however it shall be understood that frequency of operation is matter of engineering choice and not a limitation of the one or more embodiments. 
     Similarly, group delay compensation for the duplexer TX to ANT passband is provided by FGDPD module  136  by employing an all-pass IIR filter to generate group delay compensation on the combined multi carrier signal  134 . Similarly, FGDPD module  136  can be moved elsewhere in the FPGA  90  or distributed among DUC  100 - 1  to  4  chains if required. 
     As depicted in  FIG. 2B , the FGDPD  136  has an infinite impulse response all-pass filter structure and may be represented in its second order section. The FGDPD  136  takes complex weighted input samples x(n) and has complex weighted output samples y(n) with a transfer function H(z) such that y(n) is equal to x(n) convolved with H(z) following the equation below. Where
         g is a compensating complex scalar gain value,   L is the number of taps (5 in this example),   a, b are the complex weighted coefficients (poles and zeros), and   z is a delay by a sample of k (k=1 to L).       

     
       
         
           
             
               H 
               ⁡ 
               
                 ( 
                 z 
                 ) 
               
             
             = 
             
               g 
               ⁢ 
               
                 
                   ∏ 
                   
                     k 
                     = 
                     1 
                   
                   L 
                 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     1 
                     + 
                     
                       
                         b 
                         
                           1 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           k 
                         
                       
                       ⁢ 
                       
                         z 
                         
                           - 
                           1 
                         
                       
                     
                     + 
                     
                       
                         b 
                         
                           2 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           k 
                         
                       
                       ⁢ 
                       
                         z 
                         
                           - 
                           2 
                         
                       
                     
                   
                   
                     1 
                     + 
                     
                       
                         a 
                         
                           1 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           k 
                         
                       
                       ⁢ 
                       
                         z 
                         
                           - 
                           1 
                         
                       
                     
                     + 
                     
                       
                         a 
                         
                           2 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           k 
                         
                       
                       ⁢ 
                       
                         z 
                         
                           - 
                           2 
                         
                       
                     
                   
                 
               
             
           
         
       
     
     As depicted in  FIG. 2B , the input X(n)  134  is coupled to the input of amplifier  210 . The output of amplifier  210  is coupled to a path of adders  212   a ,  214   a , and so on to include  212   n  and  214   n . The output of adder  212   a  is also fed to Z −1  unit  228   a  which in turn outputs a signal to both Z −1  unit  230   a  and to a first path comprising multiplier  224   a  which multiplies the signal with coefficient −a( 1 ,k)  215   a  and feds the product into adder  222   a . The output of adder  222   a  is coupled to adder  212   a . The Z −1  unit  228   a  also provides an output to multiplier  226   a  which multiplies the signal with coefficient b( 1 ,k)  227   a  which in turn provides an output to adder  216   a . The Z −1  unit  230   a  outputs a signal to a path comprising a multiplier  220   a  which multiplies the signal with coefficient −a( 2 ,k)  221   a  and feds the output to adder  222   a . The Z −1  unit  230   a  also outputs a signal to a path comprising a multiplier  218   a  which multiplies the signal with coefficient b( 2 ,k)  219   a  and feds the output to adder  216   a . The output of adder  216   a  is coupled to adder  214   a.    
     The circuit may be replicated in one or more embodiments. In the last stage, the output of adder  212   n  is also fed to Z −1  unit  228   n  which in turn outputs a signal to both Z −1  unit  230   n  and to a path comprising multiplier  224   n  which multiplies the signal with coefficient −a( 1 ,L)  215   n  and feds the product into adder  222   n . The output of adder  222   n  is coupled to adder  212   n . The Z −1  unit  228   n  also provides an output to multiplier  226   n  which multiplies the signal with coefficient b( 1 ,L)  227   n  which in turn provides an output to adder  216   n . Z −1  unit  230   n  outputs a signal to a path comprising a multiplier  220   n  which multiplies the signal with coefficient −a( 2 ,L)  215   n  and feds the output to adder  222   n . The Z −1  unit  230   n  also outputs a signal to a path comprising a multiplier  218   a  which multiplies the signal with coefficient b( 2 ,L) and feds the output to adder  216   n . The output of adder  216   n  is coupled to adder  214   n.    
     The effectiveness of the compensation can be found in the table below. 
     CDMA IS-95 carrier FC=862.90 MHz 
     
       
         
           
               
               
               
             
               
                   
                   
               
               
                   
                 Rho (″ρ″) 
                 Compensation 
               
               
                   
                   
               
             
            
               
                   
                 0.985 to 0.990 
                 All compensation ON 
               
               
                   
                 0.975 to 0.985 
                 Filter pass band OFF; group delay ON 
               
               
                   
                 0.968 to 0.975 
                 Filter pass band ON; group delay OFF 
               
               
                   
                 0.950 to 0.970 
                 Filter pass band OFF; group delay OFF 
               
               
                   
                   
               
            
           
         
       
     
     The filter passband predistorter compensation  112  can be observed on the output spectrum, but group delay compensation  136  effect can be quantified by signal quality—Rho (“ρ”).  FIG. 6  is a representation of simulated amplitude and group delay characteristic low frequency detail of a narrow band-pass filter (Cellular  850 ).  FIG. 7  is a simulation of the FPPD effect on the output spectrum flatness. 
     The present invention has been described primarily as structures and methods for employing digital predistortion to improve passband filter performance. The description is not intended to limit the invention to the form disclosed herein. Accordingly, variants and modifications consistent with the following teachings, skill, and knowledge of the relevant art, are within the scope of the present invention. The embodiments described herein are further intended to explain modes known for practicing the invention disclosed herewith and to enable others skilled in the art to utilize the invention in equivalent, or alternative embodiments and with various modifications considered necessary by the particular application(s) or use(s) of the present invention. 
     REFERENCE LABEL LIST 
     
         
         Item Description 
           10  Remote radio head (RRH) 
           50  I/O Unit 
           60  Microcontroller Unit (MCU) 
           70  Read Only Memory ROM (non volatile memory-flash memory) 
           80  Dynamic random access memory SDRAM 
           90  Field programmable gate array (FPGA) or semi custom ASIC 
           100  Digital Up Converter (DUC) one or more −1; −2; −n 
           102  DUC input port 
           104  CPRI I/O filtering and processing 
           106  Output of  104   
           108  Channel Filtering+pulse shaping module 
           110  Output of  108   
           112  (FPPD) Filter Passband Predistorter 
           114  Output of  112   
           116  Interpolation module 
           118  Output of  116   
           120  Numerically controlled (NCO) up conversion 
           126  DUC(s) output ports 
           128  DUC to multicarrier summer 
           132  Multicarrier summer—signal generation 
           134  Output of  132   
           136  FGDPD—Filter group delay predistortion 
           138  Output of  136   
           140  CFR Crest Factor Reduction and DPD digital predistortion 
           142  Output of  140   
           300  RF transceiver ( 300 ). 
           302  TX module (or chain) includes observation path for DPD ( 140 ) 
           306  Duplexer 
           308  Transmit—receive antenna 
           310  Receive Chain 
           144  Digital output (ADC) from Receive Chain ( 310 ) to FPGA 
           146  Receive FPGA block(s) 
           202   a - n  Z −1  unit 
           204   a - n  Multiplier 
           206  Adder 
           208   a - n  Coefficients 
           210  Amplifier 
           212   a - n  Adder 
           214   a - n  Adder 
           215   a - n  Coefficients 
           216   a - n  Adder 
           218   a - n  Multiplier 
           219   a - n  Coefficients 
           220   a - n  Multiplier 
           221   a - n  Coefficients 
           222   a - n  Adder 
           224   a - n  Multiplier 
           226   a - n  Multiplier 
           227   a - n  Coefficients 
           228   a - n  Z −1  unit 
           230   a - n  Z −1  unit