Patent Publication Number: US-7591689-B2

Title: Electrical connector with improved crosstalk compensation

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a divisional of U.S. application Ser. No. 11/926,987, filed Oct. 29, 2007, now U.S. Pat. No. 7,481,681 which is a divisional of U.S. application Ser. No. 11/693,256, filed Mar. 29, 2007, now U.S. Pat. No. 7,384,315, which is a continuation of U.S. application Ser. No. 11/464,335, filed Aug. 14, 2006, now U.S. Pat. No. 7,309,261, which is a continuation of U.S. application Ser. No. 11/099,110, filed Apr. 5, 2005, now U.S. Pat. No. 7,153,168 which claims priority to U.S. Application Ser. No. 60/559,846, filed Apr. 6, 2004. All of the previous applications are herein incorporated by reference in their entireties. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates generally to electrical connectors, and more particularly, to a modular communication jack design with crosstalk compensation that is less susceptible to propagation delay effects at high frequencies. 
     BACKGROUND OF THE INVENTION 
     In the communications industry, as data transmission rates have steadily increased, crosstalk due to capacitive and inductive couplings among the closely spaced parallel conductors within the jack and/or plug has become increasingly problematic. Modular connectors with improved crosstalk performance have been designed to meet the increasingly demanding standards. Many of these improved connectors have included concepts disclosed in U.S. Pat. No. 5,997,358, the entirety of which is incorporated by reference herein. In particular, recent connectors have introduced predetermined amounts of crosstalk compensation to cancel offending near end crosstalk (NEXT). Two or more stages of compensation are used to account for phase shifts from propagation delay resulting from the distance between the compensation zone and the plug/jack interface. As a result, the magnitude and phase of the offending crosstalk is offset by the compensation, which, in aggregate, has an equal magnitude, but opposite phase. 
     Recent transmission rates, including those in excess of 500 MHz, have exceeded the capabilities of the techniques disclosed in the &#39;358 patent. Thus, improved compensation techniques are needed. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         FIG. 1  is an exploded perspective view of a communications connector, including a plug and jack; 
         FIG. 2  is a simplified schematic diagram illustrating parts of a connector assembly that are primarily responsible for causing and compensating for near end crosstalk; 
         FIG. 3  is a schematic vector diagram illustrating vectors A, B, and C on a time axis; 
         FIG. 4  is a schematic vector diagram illustrating magnitude and phase components for vectors A, B, and C on a polar axis, with reference to crosstalk vector A. 
         FIG. 5  is a schematic vector diagram illustrating vectors A, B, and C on a polar axis, with reference to compensation vector B; 
         FIGS. 6A-6C  are schematic vector polar diagrams illustrating the effect on |A+C| relative to |B| as frequency increases for a typical communications connector; 
         FIG. 7  is a graph of near end crosstalk versus frequency, illustrating crosstalk performance of a typical Cat. 6 communications connector in relation to TIA-568B requirements; 
         FIGS. 8A-8C  are schematic vector polar diagrams illustrating the effect on |A+C+D| relative to |B| as frequency increases, for a communications connector employing an embodiment of the invention; 
         FIGS. 9A-9C  are schematic vector polar diagrams illustrating the effect on |A+C| relative to |B| as frequency increases, for a communications connector employing an embodiment of the invention; 
         FIGS. 10A-10C  are schematic vector polar diagrams illustrating the effect on |A+C| relative to |B| as frequency increases, for a communications connector employing an embodiment of the invention; 
         FIGS. 11A-11C  are schematic diagrams, including equivalent circuit representations, illustrating a first embodiment of the invention; 
         FIG. 12  is a schematic diagram illustrating an alternative implementation of the first embodiment shown in  FIGS. 11A-11C ; 
         FIGS. 13A-13C  are simplified schematic diagrams illustrating a back-rotated contact design, a front-rotated contact design, and a corresponding equivalent circuit representation illustrating an embodiment of the invention; 
         FIGS. 14A and 14B  are partial perspective view diagrams illustrating front-rotated and back-rotated contact designs, respectively, in accordance with an embodiment of the invention; 
         FIG. 14C  is a partial perspective view diagram illustrating an alternative front-rotated design in accordance with an embodiment of the invention; 
         FIG. 15  a graph of near end crosstalk versus frequency, illustrating crosstalk performance of a communications connector according to an embodiment of the invention, in relation to TIA-568B requirements; 
         FIG. 16  is a right-side view illustrating a front-rotated contact configuration in a communications jack, in accordance with an embodiment of the invention; 
         FIG. 17  is a right-side view illustrating a front-rotated contact configuration in a communications jack, in accordance with another embodiment of the invention; 
         FIG. 18  is an upper right-side exploded perspective view of a connector jack in accordance with an embodiment of the present invention; 
         FIG. 19  is an upper right-side perspective view of a six-position flexible PCB in accordance with an embodiment of the invention; 
         FIG. 20  is an upper right-side perspective view of a front sled with plug interface contacts and an upward-folded flexible PCB in accordance with an embodiment of the invention; 
         FIG. 21  is an upper right-side perspective view of a front sled with plug interface contacts and a downward-folded flexible PCB in accordance with an embodiment of the invention; 
         FIG. 22  is a partial upper right-side perspective view illustrating an upward-folded flexible PCB attached to plug interface contacts in accordance with an embodiment of the invention; 
         FIG. 23  is a simplified right-side cross-sectional view of a portion of a communications connector showing arrangement of an upward-folded flexible PCB; 
         FIG. 24  is a simplified right-side cross-sectional view of a portion of a communications connector showing arrangement of a downward-folded flexible PCB; 
         FIG. 24A  is a simplified right-side cross-sectional view of a portion of a communications connector showing an alternative arrangement of a flexible PCB; 
         FIG. 25A  is an upper right-side perspective view of one embodiment of a flexible PCB that may be utilized in accordance with the present invention; 
         FIG. 25B  is a side view of one embodiment of a flexible PCB that may be utilized in accordance with the present invention; 
         FIG. 25C  is a front elevational view of one embodiment of a flexible PCB that may be utilized in accordance with the present invention; 
         FIG. 25D  is a front elevational view of a flexible PCB with the fingers in an unbent configuration, for ease of illustration, in accordance with an embodiment of the present invention; 
         FIG. 25E  is a cross-sectional view of the capacitive plates and leads in a flexible PCB in accordance with an embodiment of the present invention; 
         FIG. 25F  is a front view of a first lead and capacitive plate in a flexible PCB with the fingers in an unbent configuration, in accordance with an embodiment of the present invention; 
         FIG. 25G  is a front view of a second lead and capacitive plate in a flexible PCB with the fingers in an unbent configuration, in accordance with an embodiment of the present invention; 
         FIG. 25H  is a front view of a third lead and capacitive plate in a flexible PCB with the fingers in an unbent configuration, in accordance with an embodiment of the present invention; 
         FIG. 25I  is a front view of a fourth lead and capacitive plate in a flexible PCB with the fingers in an unbent configuration, in accordance with an embodiment of the present invention; 
         FIG. 26  is an upper right-side exploded perspective view of a connector jack employing a flexible PCB in accordance with an embodiment of the invention; 
         FIG. 27  is an upper right-side perspective view of an assembled jack in accordance with an embodiment of the invention; 
         FIG. 28  is an upper right-side perspective exploded view of a jack in accordance with an embodiment of the invention; 
         FIG. 29  is an upper right-side perspective view of a plug interface contact sub-assembly and PCB designed to accommodate 8-position plugs or 6-position plugs; 
         FIG. 30  is simplified pictorial representation of an attachment of a ferrite material structure that serves as an inductor; 
         FIG. 31  is simplified pictorial representation of two traces altered to increase coupling; 
         FIG. 32  is simplified pictorial representation of two sets of traces, one utilizing a magnetic coupler and the other utilizing magnetic material placed in through-holes; 
         FIG. 33  is simplified pictorial representation of two parallel traces on separate layers of a PCB; and 
         FIG. 34  is simplified pictorial representation of traces on a PCB with an overlay of magnetic material. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 1  is an exploded perspective illustration of a communication connector  100  comprising a plug  102  and a jack  104 , into which the plug  102  may be inserted. The plug  102  terminates a length of twisted pair communication cable (not shown), while the jack  104  may be connected to another piece of twisted pair communication cable or punch-down block (neither of which is shown in  FIG. 1 ) 
     As shown from left to right, the jack  104  includes a main housing  106  and a bottom front sled  108  and top front sled  110  arranged to support eight plug interface contacts  112 . The plug interface contacts  112  engage a PCB (Printed Circuit Board)  114  from the front via through-holes in the PCB  114 . As illustrated, eight IDCs (Insulation Displacement Contacts)  116  engage the PCB  114  from the rear via additional through-holes in the PCB  114 . A rear housing  118  having passageways for the IDCs  116  serves to provide an interface to a twisted pair communication cable or punch-down block. The general connector  100  illustrated in  FIG. 1  serves as background for the following discussion of improvements that may be made to the connector  100  to improve crosstalk performance. 
     The simplified schematic diagram of  FIG. 2  conceptually illustrates parts of a connector assembly  300  that are primarily responsible for causing near end crosstalk, as well as those that may be used to compensate for near end crosstalk. The plug  302  and plug interface contacts  304  contribute respective capacitive and inductive crosstalk components C plug +L plug  and C contacts +L contacts , which may be approximated as a lumped crosstalk vector A (see  FIG. 4 ). A compensation zone  306  on the PCB  308  provides crosstalk compensation to produce compensation vector B. To account for the phase shift of B with respect to A that will occur due to propagation delay, a near end crosstalk zone  310  (shown opposite the PCB  308  from IDCs  312 ) may contribute some additional crosstalk C to reduce the phase shift&#39;s effect on combined crosstalk. 
       FIG. 3  illustrates vectors A, B, and C on a time axis. Note that the crosstalk vectors A and C are opposite in polarity from compensation vector B. The vectors&#39; relative displacement along the time axis is caused by the physical distance of the compensation zone  306  and the crosstalk zone  310  from where the plug  302  meets the plug interface contacts  304  (causing propagation delays T 1  and T 2 ) and the relative permittivity of the intervening conduction paths. 
       FIG. 4  illustrates vectors A, B, and C on a polar axis, wherein displacement along the time axis of  FIG. 3  has been translated to phase shift with reference to crosstalk vector A. As frequency increases, the phase shift of B will grow toward A and that of C will grow in opposition to A. For relatively small phase shifts, combined crosstalk can be minimized by designing the compensation zone and crosstalk zone so that |B+C| is approximately equal to |A| at a desired null frequency. 
     For frequencies up to about 300 MHz, the multi-zone crosstalk compensation technique illustrated in  FIGS. 2-4  is suitable to comply with Cat. 6 (TIA-568B) requirements for near end crosstalk. At higher frequencies, however, this technique is unsatisfactory. To illustrate,  FIG. 5  shows vectors A, B, and C on a polar axis, but with reference to compensation vector B. To minimize combined crosstalk, |B| should be selected to be close to |A+C|. However, as frequency increases, A and C experience larger phase shifts, evidenced by larger angles from vertical on the polar axis of  FIG. 5 . Because the cosines of these increasing angles will decrease, |A+C| will become considerably less than |B|, resulting in unsatisfactory connector performance. This effect is illustrated in  FIGS. 6A-6C , where |A+C| becomes relatively smaller than |B| as frequency increases. 
       FIG. 7  shows combined crosstalk performance of a typical Cat. 6 connector using the technique discussed with reference to  FIGS. 2-6 . Note the frequency at which the NEXT crosses the TIA-568B requirements limit. 
     To improve the NEXT performance to be suitable beyond the frequencies that are feasible with the above technique, an additional coupling having a magnitude that grows disproportionate to frequency relative to a typical coupling may be included in the connector. Alternatively, one of the existing couplings can be modified to have a magnitude that varies disproportionally relative to the other couplings. Past typical connector couplings have been capacitive or mutually inductive, resulting in a magnitude that is proportional to frequency (approximately 20 dB per decade). The relative magnitudes of these typical connector couplings have remained approximately the same throughout various frequencies. By introducing a coupling that grows disproportionally relative to other couplings, the compensation for phase shifts caused by propagation delay (see  FIGS. 2-6C , above) will retard the growth of the combined crosstalk through higher frequencies. 
       FIGS. 8A-15  and their accompanying descriptions show alternative implementations of additional couplings having a magnitude that grows at a disproportionate rate relative to typical couplings, in response to frequency. Other implementations may also be used without departing from the spirit and scope of the present invention.  FIGS. 8A-10C  are vector diagrams showing desired coupling characteristics. The description of  FIGS. 8A-10C  is followed by a discussion of alternative methods for achieving the desired coupling characteristics. 
     According to a first implementation, the additional coupling is a fourth coupling, D, having a magnitude with a frequency dependency that is different than that of A, B, and C. For example, at low frequencies, A, B, and C change at a rate of 20 dB per decade, while D could change at a lower rate, such as approximately 5 dB per decade. Then, at higher frequencies (such as those greater than a null frequency of interest), D could change at a higher rate (such as 30 dB per decade), while A, B, and C remain relatively constant at 20 dB per decade. By selecting |B|−|D| to be equal to |A|+|C| at the null frequency, the combined crosstalk is near zero at low frequency, as shown by  FIG. 8A .  FIG. 8B  shows that as frequency increases, the phase angles of A and C increase, resulting in smaller vertical magnitude components to offset |B|. However, the more rapidly growing |D| increases to compensate for decreasing |A+C|.  FIG. 8C  illustrates this effect at an even higher relative frequency. 
     In a second implementation, illustrated in  FIGS. 9A-9C , compensation zone vector B is designed to have a magnitude with a frequency dependency that differs from that of A and C. For example, at low frequencies, if A and C change at a rate of 20 dB per decade, then B could be selected to vary at a lower rate, such as 15 dB per decade. At higher frequencies (such as those greater than a null frequency of interest), B could negatively change at a higher rate (such as −30 dB per decade), while A, and C remain relatively constant at 20 dB per decade. In contrast to the first implementation illustrated in  FIGS. 8A-8C , no additional coupling is needed in this second implementation. By selecting |B| to be close to |A|+|C| at the null frequency, the combined crosstalk is near zero at low frequency. As frequency increases, |A| and |C| will grow disproportionately faster than |B|, so that |B| will be close to |A+C| at increased frequencies (see  FIGS. 9B and 9C ). 
     In a third implementation, illustrated in  FIGS. 10A-10C , couplings A and C are selected to have a greater magnitude dependence on frequency than B at frequencies higher than the null frequency. For example, at low frequencies, A, B, and C could all change at a rate of 20 dB per decade. At high frequencies, however, A and C could be selected to vary at a higher rate, such as 25 dB per decade, while B remains at approximately 20 dB per decade. By selecting |B| to be close to |A|+|C| at the null frequency of interest, the combined crosstalk is near zero at low frequency. Due to the higher frequency dependencies of |A| and |C|, the more rapidly growing |A| and |C| can compensate for the decreasing |A+C| that would normally occur with increased phase angles caused by high frequency operation. Thus, low combined crosstalk can be maintained over a wider frequency range, as shown in  FIGS. 10A-10C . Of course, to vary A would likely require a change to the plug itself, which may be unacceptable in some cases. However, changing even C alone would provide some benefit. 
     The three implementations described above are merely examples of possible implementations. The relative rates of change in magnitude given in dB per decade may vary from one application to the next, depending on the specific construction and materials of the connector assembly. In addition, the concept of relative magnitude variation over frequency may be applied to improve performance at frequencies other than at or around the null frequency. The null frequency was chosen for the above examples because it serves as a good starting point for making adjustments to improve high frequency operation. For current communications applications, null frequencies are generally observed around 100-250 MHz. Different connector designs will likely exhibit different null frequencies. 
     In a preferred embodiment, the communication jack includes plug interface contacts for making electrical contact with the plug contacts in a plug, where the plug interface contacts and plug contacts introduce crosstalk to the connector. The crosstalk has an associated first frequency dependency based on a frequency of a communication signal being communicated. The jack has at least two crosstalk compensation zones, with at least one of the crosstalk compensation zones including a coupling having an associated second frequency dependency that substantially differs from the first frequency dependency associated with the plug interface contacts and plug contacts. The first frequency dependency is a magnitude change of approximately 20 dB per decade. The second frequency dependency is a magnitude that changes from approximately 0 dB per decade at a first frequency to approximately 20 dB per decade at a second frequency. In a second preferred embodiment, the second frequency dependency is a magnitude that changes from approximately 20 dB per decade at a first frequency to less than 20 dB per decade at a second frequency. Finally, in a third preferred embodiment, the second frequency dependency is a magnitude change of 20 dB per decade, and the first frequency dependency is a magnitude that changes from approximately 20 dB per decade at a first frequency to greater than 25 dB per decade at a second frequency. 
     The adjustments to magnitude dependency on frequency may be made using several alternative techniques. The following discussion sets forth five of these techniques; however, others may be used without departing from the spirit and scope of the present invention. 
     Coupling alternative #1:  FIGS. 11A-11C  illustrate an example of a first embodiment, in which a capacitance is placed in series with a mutual-inductive coupling. The mutual inductive coupling generates a current in the reverse direction of the current flowing through the capacitor, as shown in  FIG. 11A , self inductance equivalent circuit  11 B, and impedance equivalent circuit  11 C. At low frequencies, coupling through the capacitor is low; therefore, the reverse current generated in the secondary side of the inductance is also low. With rising frequency, coupling through the capacitor will rise, increasing the current through the primary side of the inductor, thereby causing a higher reverse current through the secondary side of the inductor. As a result, coupling declines proportionally to frequency. In a preferred embodiment, the “balanced source”  1262  shown in  FIG. 11A  is pairs  3  and  6 , while the “balanced sink”  1264  is pairs  4  and  5 .  FIG. 12  shows an alternative arrangement of this embodiment, with pairs  3 - 4  and  5 - 6  illustrated on the left side. 
       FIGS. 13A-13C  illustrate how coupling alternative #1 may be implemented in either a back-rotated plug contact design  1300  or a front-rotated plug contact design  1302 . An example showing the resulting couplings in the case where compensation capacitance is implemented on an interface PCB  1304  is illustrated in the simplified equivalent circuit  1306 . 
       FIGS. 14A and 14B  illustrate the location in a front-rotated design  1400  and a back-rotated design  1406  where the capacitive couplings may be located. In the front-rotated design  1400 , the capacitance is placed in the tip nose region  1404  in a way that avoids physical interference with the plug  1402 . In the back-rotated design  1406 , the capacitance may again be located in the tip nose region  1410 , which is on the opposite side of the plug  1408  when compared to the front-rotated design. For the back-rotated design  1406 , the capacitance may be placed above or below the contacts of the tip nose region  1410 , so long as it does not physically interfere with insertion of the plug  1408 . The placements shown in  FIGS. 14A and 14B  result in capacitive couplings C 35  and C 46  (from pairs  3  and  5  and  4  and  6 , respectively) and mutual inductive couplings M 43  and M 56  (from pairs  4  and  3  and  5  and  6 , respectively). 
       FIG. 14C  illustrates another location in an alternative front-rotated design  1412 , as schematically illustrated in  FIG. 12 , where the couplings may be located. In the alternative front-rotated design  1412 , the couplings are placed even closer to the point of electrical contact between the plug  1414  and the plug interface contacts  1416 . This closer placement results from locating the couplings on the opposite side of the plug interface contacts  1416  from the plug  1414 . This is achieved by moving the inductive compensation from the conductors seen in the Tip Nose  1404  of  FIG. 14A  into a PCB, such as the flexible PCB shown in  FIG. 24A . This results in reduced propagation delay and thus, reduced phase shift, which in turn provides better crosstalk performance. 
     Coupling alternative #2: In a second alternative, the coupling takes the form of a capacitance that varies with frequency relative to other couplings. One example of such a capacitance is a capacitor having a dielectric with a permittivity that changes with frequency. 
     Coupling alternative #3: According to a third alternative, the coupling is mutually inductive with a relative frequency-dependent inductance. One example of such an inductance is an inductive element composed of a ferrite material. Ferrites (e.g. compounds with iron oxide and nickel-zinc or manganese-zinc) typically exhibit permeabilities that vary greatly as a function of frequency starting at frequencies of around 100 kHz to 1 GHz. For example, a mixture of iron oxide and nickel-zinc has an initial permeability ranging from 10 to 1,500 over a range of 1 MHz to 1 GHz. 
     Coupling alternative #4: In a fourth alternative, the coupling is a capacitance in series with one or more resistors that are frequency-dependent. For example, a conductor or semiconductor resistor can be constructed to take advantage of the skin-effect to increase resistance at high frequencies. 
     Coupling alternative #5: According to a fifth alternative, a capacitance is placed in series with a self-inductive coupling. Increased inductance at higher frequencies will result in less coupling through the capacitance. 
       FIG. 15  shows improved combined crosstalk performance of a typical Cat. 6 connector that may be obtained using the inventive techniques discussed above with reference to  FIGS. 8A-14 . Note that the frequency at which the NEXT crosses the TIA-568B requirements limit is much higher than in  FIG. 7 . 
     The high frequency effects described with reference to  FIGS. 2-7 , and the need to implement the above solutions to achieve acceptable high-frequency operation, arise primarily from the physical distance between the plug interface contacts and first compensation. By decreasing this distance, better performance (i.e. less phase shift due to propagation delay) may be attained at high frequencies. For example, moving the first compensation point to a point less than approximately 0.200 inches from the plug/jack interface provides better crosstalk performance.  FIGS. 16-28  illustrate physical changes that may be made to a jack to shorten the distance between the plug interface contacts and first compensation. These changes may be made in lieu of, or in combination with, the techniques described above. Optimal crosstalk performance will result from implementing the combination. 
       FIG. 16  is a right-side schematic diagram illustrating a front rotated contact configuration  1600 , including a plurality of plug interface contacts  1602  disposed in a contact carrier and front sled  1604  and a vertical interface PCB  1606  having a contact portion  1608  connected to a crosstalk compensation zone (not shown). Compared to typical plug interface contacts, the plug interface contacts  1602  are longer so that they come into contact with the contact portion  1608  of the vertical interface PCB  1606 . As a result, the distance  1610  between the contact portion  1608  and the point at which contact is made between an inserted plug and the plug interface contacts  1602  is significantly smaller than for typical plug interface contacts, as can be seen by comparing the distance  1610  to distance  1700  in  FIG. 17 . Because the improved design has a shorter distance between the plug contact and the first compensation, propagation delay is lessened, resulting in a smaller phase shift. This, in turn, enables better crosstalk compensation and operation at higher frequencies than would be possible without such a design. It should be noted that  FIG. 17  includes inductive couplings shown generally at  1702 , which assist in crosstalk compensation. 
       FIG. 18  is an upper right-side exploded perspective view of a connector jack  1800  employing the above concept. The jack  1800  includes a bottom front sled  1804  and a top front sled  1808 , each mechanically attached to a plurality of plug interface contacts  1806 . A first end  1810  of the plug interface contacts  1806  may be inserted into through-holes in an interface PCB  1812 , while a second end  1814  includes plug interface contact ends that are longer than for a typical jack to allow contact with a compensation zone on the interface PCB  1812 . The sub-assembly comprising the bottom front sled  1804 , plug interface contacts  1806 , top front sled  1808 , and interface PCB  1812  is then inserted into a housing  1802 . Also to be inserted into through-holes on the interface PCB  1812  are a plurality of IDCs  1816 . A rear sled  1820  is snapped into the housing  1802 . A wire containment cap  1818  is configured to accept a four-pair twisted-pair communication cable for connection to the IDCs  1816  through the rear sled  1820 . The wire containment cap  1818  may then be snapped onto the rear sled  1820 , forming an integrated communication jack assembly. 
     While the above technique uses an alternative conductor path between the plug interface contacts and the first compensation, a second technique consists of placing the first compensation zone closer to the plug contact point by attaching a flexible PCB to the plug interface contacts. As an example, pad capacitors could be etched onto the flexible PCB to provide capacitive crosstalk compensation, thereby improving the electrical performance of the jack. 
       FIG. 19  shows a six-position flexible PCB  1900  having six fingers  1902  that may be used to attach the flexible PCB  1900  to plug interface contacts  2000  carried in a front sled  2002 , as shown in  FIG. 20 . While a six-position flexible PCB  1900  is shown, an eight-position implementation is also possible. A six-position design may be preferred to avoid damage to standard RJ-45 jacks when a six-position RJ-45 plug is inserted. A standard six-position RJ-45 plug has plastic that protrudes further than the six contacts, which may lead to excessive displacement of plug interface contacts in the jack. The six-position flexible PCB  1900  allows plug interface contacts  1  and  8  to be displaced further than plug interface contacts  2  though  7 . The flexible PCB  1900  is preferably constructed of a layer of copper adhered to a polyester or polyamide substrate. The copper can be removed (e.g. by etching) in various configurations to create a crosstalk compensation zone. The fingers  1902  of the flexible PCB  1900  may be attached to the plug interface contacts  2000  in any of a number of ways. Attachment techniques may include ultrasonically welding or heat soldering, for example. 
       FIGS. 21 and 22  are perspective illustrations showing that the flexible PCB  1900  may be folded upward or downward. Other orientations and configurations are also possible.  FIG. 22  also shows a suitable region of the plug interface contacts  2000  for attaching the fingers  1902  to the plug interface contacts  2000 . Depending on the number of fingers  1902 , the flexible PCB  1900  will be attached to the appropriate contacts for tuning. 
       FIGS. 23 and 24  are simplified right-side cross-sectional views illustrating that the flexible PCB  1900  may experience deflection upward ( FIG. 23 ) or downward ( FIG. 24 ) in the jack as the plug interface contacts travel in response to insertion of a plug. As the plug is inserted into the jack, the flexible PCB  1900  follows the free deflection of each contact regardless of whether or not it is attached to the flexible PCB  1900 . The fingers of the flexible PCB  1900  also accommodate the natural variation in contact deflection due to variation in the plug contact termination height. Clearance may need to be built into the housing for the upward-deflecting flexible PCB  1900  of  FIG. 23  or into the front top sled for the downward-deflecting flexible PCB  1900  of  FIG. 24 . Note that the vertically-spaced layout of plug interface contacts  2350  shown in  FIGS. 23  and  24  advantageously provides additional inductive crosstalk compensation. While this layout is preferred, other layouts may alternatively be used. 
       FIG. 24A  is a simplified right-side cross-sectional view illustrating an alternative placement of the flexible PCB  1900  on the plug interface contacts  2350 . In this alternative placement, which may, for example utilize the design shown in  FIG. 14C , the flexible PCB  1900  and plug (not shown) are on opposite sides of the plug interface contacts  2350 . This allows the couplings on the flexible PCB  1900  to be very close to the plug contact point  2370 , resulting in reduced propagation delay and thus, reduced phase shift. This, in turn, provides better crosstalk performance. To allow for deflection of the plug interface contacts  2350  when a plug is inserted, the flexible PCB  1900  may be designed to avoid contact with other parts of the jack, such as the lower part of the plug interface contacts  2350 . 
       FIG. 25A  is an upper right-side perspective view,  FIG. 25B  is a side view, and  FIG. 25C  is a front elevational view of one embodiment of a flexible PCB  2400  that may be utilized in accordance with the present invention to provide crosstalk compensation. The PCB  2400  includes a main portion  2402  and attachment fingers, such as the finger  2404 . The main portion  2402  supports a plurality of capacitive plates (in this case, four plates, corresponding to plug interface contacts  3 - 6 ) to provide capacitive coupling. As will be illustrated in  FIGS. 25D-I , the leads to the capacitive plates provide an inductive coupling component as well. The fingers  2404  serve as an attachment mechanism for attaching the PCB  2400  to the plug interface contacts, using one of the schemes shown in  FIGS. 23-24A , for example. While any suitable attachment technique may be used, in the illustrated embodiment, a resistance weld rivet  2406  is used. In addition to attaching the PCB  2400  to the plug interface contacts (or another conductor connected to the plug interface contacts), the rivet  2406  acts as a contact post for the capacitive plates and their leads. This is illustrated in  FIGS. 25B-I , which show four layers of capacitive plates  2412  and leads ( 2408   a - d ), through which the rivet  2406  protrudes to make appropriate contact in the fingers  2404 . 
       FIG. 25D  is a front elevational view of the PCB  2400  with the fingers in an unbent configuration, for ease of illustration.  FIG. 25E  is a cross-sectional view of the capacitive plates and leads as viewed upward from the bottom of the PCB  2400  toward line A/A in  FIG. 25D . Note that  FIG. 25E  does not show portions of the PCB  2400  that merely support the capacitive plates and leads or serve as a dielectric or insulator.  FIGS. 25D-I  show how the capacitive plates and leads are placed with respect to one another to result in a relatively high density of inductive coupling in a relatively short distance. For example, in  FIG. 25D , the capacitive plate  2412   a  and lead  2408   a  for conductor  5  is the topmost plate and lead shown, having a sideways “U” shape. The same “U” shape, but with varying orientation, is used for conductors  3 ,  4 , and  6 , as shown by the dashed and solid lines of  FIG. 25D . The physical placement and overlapping area of the capacitive plates determines the amount of capacitive coupling. Similarly, the separation of the leads from one another and the length of overlap determine the amount of inductive coupling.  FIG. 25E  also illustrates the relative direction of current flow in the respective leads, which provides a high density of inductive coupling.  FIGS. 25F-25I  show, respectively, leads  2408   a - d  and capacitive plates  2412   a - d  associated with, respectively, fifth, third, sixth, and fourth conductors of an eight-conductor jack. 
       FIG. 26  is an upper right-side exploded perspective view of a connector jack  2500  employing the flexible PCB concept. The jack  2500  includes a bottom front sled  2504  and a top front sled  2508 , each mechanically attached to a plurality of plug interface contacts  2506 . A first end  2510  of the plug interface contacts  2506  may be inserted into through-holes in an interface PCB  2512 , while a second end  2514  is attached to a flexible PCB  2516  that provides crosstalk compensation. The sub-assembly comprising the bottom front sled  2504 , plug interface contacts  2506 , top front sled  2508 , interface PCB  2512 , and flexible PCB  2516  is then inserted into a housing  2502 . Also to be inserted into through-holes on the interface PCB  2512  are a plurality of IDCs  2518 . A rear sled  2520  is snapped into the housing  2502 . A wire containment cap  2522  is configured to accept a four-pair twisted-pair communication cable (not shown) for connection to the IDCs  2518  through the rear sled  2520 . The wire containment cap  2522  may then be snapped onto the rear sled  2520 , forming an integrated communication jack assembly. 
     While  FIGS. 19-26  are described with reference to a flexible PCB, this is merely one embodiment, and other embodiment using rigid PCBs or other compensation schemes may also be possible without departing from the intended scope of the invention. A flexible PCB may assist in meeting mechanical constraints that may exist in some connector designs. 
     Another technique for shortening the distance between the crosstalk compensation zone and the interface between the plug and plug interface contacts will now be described with reference to  FIGS. 27-29 . In this third technique, a back-rotated plug interface contact design is used in conjunction with an underlying compensation PCB to provide crosstalk compensation extremely close to the interface between the plug and plug interface contacts. As a result, propagation delays are minimized, as is the phase shift of the crosstalk compensation. This simplifies the overall compensation scheme by reducing the number of zones of crosstalk and compensation, which allows for better operation at high frequencies. 
       FIG. 27  is an upper right-side perspective view of an assembled jack  2600 . The jack  2600  includes a housing  2602  designed to accept a plug (not shown), a rear sled  2604 , and a wire containment cap  2606  configured to accept a communication cable (not shown). The housing  2602 , rear sled  2604 , and wire containment cap  2606  latch together to form the assembled jack  2600 . 
       FIG. 28  is an upper right-side perspective exploded view of the jack  2600 . In addition to the housing  2602 , rear sled  2604 , and wire containment cap  2606  described with reference to  FIG. 27 , the jack  2600  includes a PCB support  2708  designed to support a compensation PCB  2710  and an interface PCB  2712 . A plurality of plug interface contacts  2714  have first ends  2716  pressed into through-holes in the interface PCB  2712  and second ends  2718 , at least some of which slide along the compensation PCB  2710  as a plug is received into the jack  2600 . A plurality of IDCs  2720  are inserted in through-holes in the interface PCB  2712 .  FIG. 29  shows a closer perspective view of this plug interface contact sub-assembly (with the exception of IDCs  2720 ), which is inserted into the housing  2602 , prior to the rear sled  2604  being snapped onto the housing  2602 . Assembly of the jack  2600  further includes positioning and installing a communication cable in the wire containment cap  2606 , which is then snapped onto the rear sled  2604 . 
     The plug interface contact sub-assembly (without IDCs  2720 ) shown in  FIG. 29  is designed to accommodate either 8-position plugs or 6-position plugs. When an 8-position plug is inserted into the jack, a downward force causes contacts  2  through  7  to slide along the compensation PCB  2710 . Contacts  1  and  8  deflect, but don&#39;t slide along the compensation PCB  2710 . In contrast, when a 6-position plug is inserted into the jack, contacts  2  through  7  still slide along the compensation PCB  2710 . However, contacts  1  and  8  deflect more than contacts  2  through  7 , due to additional plastic material on the 6-position plug. The clearance over the compensation PCB  2710  provided by plug interface contacts  1  and  8  allows for this additional deflection, while maintaining adequate normal force between the plug and plug interface contacts  2714 . 
     Inductance Enhancement for Compensation Circuits 
     The compensation circuits described above with reference to  FIGS. 11A-14C  may be realized using standard layout and processing techniques composed of well-known electrical components. Additionally, generating mutual inductance circuits with substantial inductive properties to act as these compensators is relatively simple, when limits are not placed on the trace length of the circuit. However, the limited space provided by the PCB board attached to the plug interface contacts within the jack housing requires novel processing techniques and devices in order to create optimal inductive properties in as short of a trace as possible. These techniques should allow phase delay to be effectively introduced into the compensation circuitry despite the shortened trace lengths required of limited PCB area. 
     One technique is to use magnetic ferrite materials to increase the mutual inductance between two signal traces. The magnetic material reacts strongly to the movement of electrical charges in a first signal trace, which also generate a magnetic flux. This magnetic flux is exhibited in the orientation of magnetic poles with the magnetic material, which then influences the moving electrical charges associated with a second electrical trace. Essentially, the magnetic material acts as a medium by which the two signal traces can be magneto-electrically coupled to a degree determined by the geometry and magnetic properties of the ferrous or magnetic material used.  FIG. 30  shows an attachment of a ferrite material structure  3000  that serves as external inductor element for the two signal traces  3002  running through it. The core structure may be in the shape of several arches with the traces passing below the structure. Alternatively, the structure may have a solid half-cylindrical shape, or may be in the form of several rectangular arches. The external magnetic core may be composed of powdered iron, iron, nickel, steel, or a composite of these metals. Alternatively, it may be composed of another magnetic ferrite material with magneto-electric inductive properties. The magnetic core may be fabricated separately from the board, and may be soldered, glued, or snapped into place at pre-fabricated sites on the PCB  3004 . Attaching this component may be performed at a different site than that of the PCB manufacturer after PCB processing has been completed. 
       FIG. 31  shows another method that can be used to increase mutual inductance between signal traces. In the method shown, no external components are required to generate the inductive coupling between the traces. Rather, the geometry of the traces themselves is altered to maximize coupling between the two signals. In this example, one trace  3100  is formed into a first winding  3102 , while the second trace  3104  is formed into a second winding  3106 . The two windings overlap by a specified amount and geometry, allowing for an increased interaction between the two traces per trace length. Alternatively, different trace geometries may be used in order to increase the inductive coupling between the signals, such as elliptical or rectangular spirals. 
       FIG. 32  illustrates two similar methods that may be used to increase the mutual inductance between signal traces. Like the first method presented, the methods shown in  FIG. 32  utilize magnetic core materials to increase the inductive coupling between two signal traces. In one method, the coupling is achieved by placing a magnetic coupler  3200  directly over two parallel traces  3202  and  3204 . The magnetic material may be applied to the surface of the board  3206  using a variety of techniques. For example, the material may be melted and deposited onto the surface using a drop dispenser, it may be screened on, it may be added using an immersion and etch process, it may be rolled on, or the magnetic materials may be added using a similar process that allows for the patterned and localized deposition of material onto the surface of the circuit board. 
     In another method shown in  FIG. 32 , the magnetic coupling material may be inserted into the PCB  3206  through fabricated holes  3208  in the board. The holes  3208  may then be filled with magnetic material  3210  using, for example, a screening process. Alternatively, the magnetic material  3210  may be press fit into the PCB. The holes  3208  may be circular with cylindrical magnetic plugs used to fill the vacancies. Alternatively, the holes may consist of a different geometry that allows for inductive coupling between the traces through the magnetic core material. 
     In both embodiments shown in  FIG. 32 , the magnetic material  3210  may be any magnetic ferrite material, such as those described above. Additionally, the magnetic components may ideally be incorporated into the PCB manufacturing process. However, the addition of the magnetic couplers may also take place after the board has been created at a different site from the PCB manufacturer. 
       FIG. 33  illustrates a similar method to the embodiments shown in  FIG. 32 . However, in this embodiment, the two signal traces  3300  and  3302  are located in parallel on separate layers within the PCB  3304 . Holes  3306  are drilled into the PCB  3304  next to the signal traces  3300  and  3302  and are then filled with magnetic material. The signal traces  3300  and  3302  may be laid out so that they wrap around the magnetic cores, thereby increasing the amount of coupling induced by the magnetic material. Alternatively, other layouts may be used that either increase or decrease the amount of coupling, as required by the electrical specifications of the circuit. Filling the holes  3306  with magnetic core material may be accomplished via a screening process. The creation of the PCB holes  3306  and subsequent filling with magnetic material may be accomplished during the PCB manufacturing process, although such processing may also take place following the creation of the board and at a different site from the PCB manufacturer. 
     Another method for increasing the mutual inductance between signal traces is illustrated in  FIG. 34 . In this method, the signal traces  3400  are fabricated onto PCB  3402  in the normal fashion. After the traces are created, an internal layer  3404  of magnetic core material is laid on top of the board followed by another capping layer of PCB material  3406 . As a result, a layer of magnetic material may be embedded within the circuit board. Alternatively, the internal layer  3404  of magnetic core material may be patterned and selectively removed prior to the application of the capping PCB layer  3406 . This would allow the magnetic material to be present only in specific areas where increased inductive coupling is desired, and would also prevent inadvertent coupling between unrelated signal traces. The creation of this type of circuit would need to be performed at the PCB manufacturer site and may require additional processing steps to incorporate the magnetic material into the board. 
     All of the above methods may be used to increase the inductive coupling per trace length in PCB manufactured circuits. These methods help to realize the crosstalk compensation circuits necessary for mitigating propagation delay effects at high frequencies in modular communication jacks. 
     Many modifications and other embodiments of the invention will come to mind to one skilled in the art to which this invention pertains having the benefit of the teachings presented in the foregoing descriptions and the associated drawings. Therefore, it is to be understood that the invention is not to be limited to the specific embodiments disclosed and that modifications and other embodiments are intended to be included within the spirit and scope of the present invention. Although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation.