Patent Publication Number: US-11050412-B2

Title: Acoustic filter using acoustic coupling

Description:
RELATED APPLICATIONS 
     This application claims the benefit of provisional patent application Ser. No. 62/385,340, filed Sep. 9, 2016, the disclosure of which is hereby incorporated herein by reference in its entirety. 
    
    
     FIELD OF THE DISCLOSURE 
     The present disclosure relates to acoustic filters that employ acoustic resonators, and in particular to an enhanced acoustic filter using acoustic coupling. 
     BACKGROUND 
     Acoustic resonators, such as Surface Acoustic Wave (SAW) resonators and Bulk Acoustic Wave (BAW) resonators, are used in many high-frequency communication applications. In particular, SAW resonators are often employed in filter networks that operate frequencies up to 1.8 GHz, and BAW resonators are often employed in filter networks that operate at frequencies above 1.5 GHz. Such filters need to have flat passbands, have steep filter skirts and squared shoulders at the upper and lower ends of the passband, and provide excellent rejection outside of the passband. SAW- and BAW-based filters also have relatively low insertion loss, tend to decrease in size as the frequency of operation increases, and are relatively stable over wide temperature ranges. 
     As such, SAW- and BAW-based filters are the filter of choice for many 3rd Generation (3G) and 4th Generation (4G) wireless devices and are destined to dominate filter applications for 5th Generation (5G) wireless devices. Most of these wireless devices support cellular, wireless fidelity (Wi-Fi), Bluetooth, and/or near field communications on the same wireless device and, as such, pose extremely challenging filtering demands. While these demands keep raising the complexity of wireless devices, there is a constant need to improve the performance of acoustic resonators and filters that are based thereon. 
     To better understand acoustic resonators and various terminology associated therewith, the following provides an overview of a BAW resonator. However, the concepts described herein may employ any type of acoustic resonator and are not limited to SAW- and BAW-based resonators. An exemplary BAW resonator  10  is illustrated in  FIG. 1 . The BAW resonator  10  generally includes a substrate  12 , a reflector  14  mounted over the substrate  12 , and a transducer  16  mounted over the reflector  14 . The transducer  16  rests on the reflector  14  and includes a piezoelectric layer  18 , which is sandwiched between a top electrode  20  and a bottom electrode  22 . The top and bottom electrodes  20  and  22  may be formed of Tungsten (W), Molybdenum (Mo), Platinum (Pt), or like material, and the piezoelectric layer  18  may be formed of Aluminum Nitride (AlN), Zinc Oxide (ZnO), or other appropriate piezoelectric material. Although shown in  FIG. 1  as each including a single layer, the piezoelectric layer  18 , the top electrode  20 , and/or the bottom electrode  22  may include multiple layers of the same material, multiple layers in which at least two layers are different materials, or multiple layers in which each layer is a different material. 
     The BAW resonator  10  is divided into an active region  24  and an outside region  26 . The active region  24  generally corresponds to the section of the BAW resonator  10  where the top and bottom electrodes  20  and  22  overlap and also includes the layers below the overlapping top and bottom electrodes  20  and  22 . The outside region  26  corresponds to the section of the BAW resonator  10  that surrounds the active region  24 . 
     For the BAW resonator  10 , applying electrical signals across the top electrode  20  and the bottom electrode  22  excites acoustic waves in the piezoelectric layer  18 . These acoustic waves primarily propagate vertically. A primary goal in BAW resonator design is to confine these vertically propagating acoustic waves in the transducer  16 . Acoustic waves traveling upward are reflected back into the transducer  16  by the air-metal boundary at the top surface of the top electrode  20 . Acoustic waves traveling downward are reflected back into the transducer  16  by the reflector  14  or by an air cavity, which is provided just below the transducer in a Film BAW Resonator (FBAR). 
     The reflector  14  is typically formed by a stack of reflector layers (RL) 28, which alternate in material composition to produce a significant reflection coefficient at the junction of adjacent reflector layers  28 . Typically, the reflector layers  28  alternate between materials having high and low acoustic impedances, such as tungsten (W) and silicon dioxide (SiO 2 ). While only five reflector layers  28  are illustrated in  FIG. 1 , the number of reflector layers  28  and the structure of the reflector  14  varies from one design to another. 
     The magnitude (Z) and phase (ϕ) of the electrical impedance as a function of the frequency for a relatively ideal BAW resonator  10  is provided in  FIG. 2 . The magnitude (Z) of the electrical impedance is illustrated by the solid line, whereas the phase (ϕ) of the electrical impedance is illustrated by the dashed line. A unique feature of the BAW resonator  10  is that it has both a resonance frequency and an anti-resonance frequency. The resonance frequency is typically referred to as the series resonance frequency (f s ), and the anti-resonance frequency is typically referred to as the parallel resonance frequency (f p ). The series resonance frequency (f s ) occurs when the magnitude of the impedance, or reactance, of the BAW resonator  10  approaches zero. The parallel resonance frequency (f p ) occurs when the magnitude of the impedance, or reactance, of the BAW resonator  10  peaks at a significantly high level. In general, the series resonance frequency (f s ) is a function of the thickness of the piezoelectric layer  18  and the mass of the bottom and top electrodes  20  and  22 . 
     For the phase, the BAW resonator  10  acts like an inductance that provides a 90° phase shift between the series resonance frequency (f s ) and the parallel resonance frequency (f p ). In contrast, the BAW resonator  10  acts like a capacitance that provides a −90° phase shift below the series resonance frequency (f s ) and above the parallel resonance frequency (f p ). The BAW resonator  10  presents a very low, near zero, resistance at the series resonance frequency (f s ) and a very high resistance at the parallel resonance frequency (f p ). The electrical nature of the BAW resonator  10  lends itself to the realization of a very high Q (quality factor) inductance over a relatively short range of frequencies, which has proven to be very beneficial in high-frequency filter networks, especially those operating at frequencies around 1.8 GHz and above. 
     Unfortunately, the phase (φ) curve of  FIG. 2  is representative of an ideal phase curve. In reality, approaching this ideal is challenging. A typical phase curve for the BAW resonator  10  of  FIG. 1  is illustrated in  FIG. 3A . Instead of being a smooth curve, the phase curve of  FIG. 3A  includes ripple below the series resonance frequency (f s ), between the series resonance frequency (f s ) and the parallel resonance frequency (f p ), and above the parallel resonance frequency (f p ). The ripple is the result of spurious modes, which are caused by spurious resonances that occur in corresponding frequencies. While the vast majority of the acoustic waves in the BAW resonator  10  propagate vertically, various boundary conditions about the transducer  16  result in the propagation of lateral (horizontal) acoustic waves, which are referred to as lateral standing waves. The presence of these lateral standing waves reduces the potential Q associated with the BAW resonator  10 . 
     As illustrated in  FIG. 4 , a border (BO) ring  30  is formed on or within the top electrode  20  to suppress certain of the spurious modes. The spurious modes that are suppressed by the BO ring  30  are those above the series resonance frequency (f s ), as highlighted by circles A and B in the phase curve of  FIG. 3B . Circle A shows a suppression of the ripple, and thus of the spurious mode, in the passband of the phase curve, which resides between the series resonance frequency (f s ) and the parallel resonance frequency (f p ). Circle B shows suppression of the ripple, and thus of the spurious modes, above the parallel resonance frequency (f p ). Notably, the spurious mode in the upper shoulder of the passband, which is just below the parallel resonance frequency f p , and the spurious modes above the passband are suppressed, as evidenced by the smooth or substantially ripple free phase curve between the series resonance frequency (f s ) and the parallel resonance frequency (f p ) and above the parallel resonance frequency (f p ). 
     The BO ring  30  corresponds to a mass loading of the portion of the top electrode  20  that extends about the periphery of the active region  24 . The BO ring  30  may correspond to a thickened portion of the top electrode  20  or the application of additional layers of an appropriate material over the top electrode  20 . The portion of the BAW resonator  10  that includes and resides below the BO ring  30  is referred to as a BO region  32 . Accordingly, the BO region  32  corresponds to an outer, perimeter portion of the active region  24  and resides inside of the active region  24 . 
     While the BO ring  30  is effective at suppressing spurious modes above the series resonance frequency (f s ), the BO ring  30  has little or no impact on those spurious modes below the series resonance frequency (f s ), as shown by the ripples in the phase curve below the series resonance frequency (f s ) in  FIG. 3B . A technique referred to as apodization is often used to suppress the spurious modes that fall below the series resonance frequency (f s ). 
     Apodization tries to avoid, or at least significantly reduce, any lateral symmetry in the BAW resonator  10 , or at least in the transducer  16  thereof. The lateral symmetry corresponds to the footprint of the transducer  16 , and avoiding the lateral symmetry corresponds to avoiding symmetry associated with the sides of the footprint. For example, one may choose a footprint that corresponds to a pentagon instead of a square or rectangle. Avoiding symmetry helps reduce the presence of lateral standing waves in the transducer  16 . Circle C of  FIG. 3C  illustrates the effect of apodization in which the spurious modes below the series resonance frequency (f s ) are suppressed, as evidence by the smooth or substantially ripple free phase curve below the series resonance frequency (f s ). Assuming no BO ring  30  is provided, one can readily see in  FIG. 3C  that apodization fails to suppress those spurious modes above the series resonance frequency (f s ). As such, the typical BAW resonator  10  employs both apodization and the BO ring  30 . 
     As noted previously, BAW resonators  10  are often used in filter networks that operate at high frequencies and require high Q values. A basic ladder network  40  is illustrated in  FIG. 5A . The ladder network  40  includes two series resonators B SER  and two shunt resonators B SH , which are arranged in a traditional ladder configuration. Typically, the series resonators B SER  have the same or similar first frequency response, and the shunt resonators B SH  have the same or similar second frequency response, which is different from the first frequency response, as shown in  FIG. 5B . In many applications, the shunt resonators B SH  are detuned versions of the series resonators B SER . As a result, the frequency responses for the series resonators B SER  and the shunt resonators B SH  are generally very similar, yet shifted relative to one another such that the parallel resonance frequency (f p,SH ) of the shunt resonators approximates the series resonance frequency (f s,SER ) of the series resonators B SER . Note that the series resonance frequency (f s,SH ) of the shunt resonators B SH  is less than the series resonance frequency (f s,SER ) of the series resonators B SER . The parallel resonance frequency (f p,SH ) of the shunt resonators B SH  is less than the parallel resonance frequency (f p,SER ) of the series resonators B SER . 
       FIG. 5C  is associated with  FIG. 5B  and illustrates the response of the ladder network  40 . The series resonance frequency (f s,SH ) of the shunt resonators B SH  corresponds to the low side of the passband&#39;s skirt (phase  2 ), and the parallel resonance frequency (f p,SER ) of the series resonators B SER  corresponds to the high side of the passband&#39;s skirt (phase  4 ). The substantially aligned series resonance frequency (f s,SER ) of the series resonators B SER  and the parallel resonance frequency (f p,SH ) of the shunt resonators B SH  fall within the passband.  FIGS. 6A through 6E  provide circuit equivalents for the five phases of the response of the ladder network  40 . During the first phase (phase  1 ,  FIGS. 5C, 6A ), the ladder network  40  functions to attenuate the input signal. As the series resonance frequency (f s,SH ) of the shunt resonators B SH  is approached, the impedance of the shunt resonators B SH  drops precipitously such that the shunt resonators B SH  essentially provide a short to ground at the series resonance frequency (f s,SH ) of the shunt resonators (phase  2 ,  FIGS. 5C, 6B ). At the series resonance frequency (f s,SH ) of the shunt resonators B SH  (phase  2 ), the input signal is essentially blocked from the output of the ladder network  40 . 
     Between the series resonance frequency (f s,SH ) of the shunt resonators B SH  and the parallel resonance frequency (f p,SER ) of the series resonators B SER , which corresponds to the passband, the input signal is passed to the output with relatively little or no attenuation (phase  3 ,  FIGS. 5C, 6C ). Within the passband, the series resonators B SER  present relatively low impedance, whereas the shunt resonators B SH  present relatively high impedance, wherein the combination of the two leads to a flat passband with steep low- and high-side skirts. As the parallel resonance frequency (f p,SER ) of the series resonators B SER  is approached, the impedance of the series resonators B SER  becomes very high, such that the series resonators B SER  essentially present themselves as open at the parallel resonance frequency (f p,SER ) of the series resonators (phase  4 ,  FIGS. 5C, 6D ). At the parallel resonance frequency (f p,SER ) of the series resonators B SER  (phase  4 ), the input signal is again essentially blocked from the output of the ladder network  40 . 
     During the final phase (phase  5 ,  FIGS. 5C, 6E ), the ladder network  40  functions to attenuate the input signal, in a similar fashion to that provided in phase  1 . As the parallel resonance frequency (f p,SER ) of the series resonators B SER  is passed, the impedance of the series resonators B SER  decreases and the impedance of the shunt resonators B SH  normalizes. Thus, the ladder network  40  functions to provide a high Q passband between the series resonance frequency (f s,SH ) of the shunt resonators B SH  and the parallel resonance frequency (f p,SER ) of the series resonators B SER . The ladder network  40  provides extremely high attenuation at both the series resonance frequency (f s,SH ) of the shunt resonators B SH  and the parallel resonance frequency (f p,SER ) of the series resonators. The ladder network  40  provides good attenuation below the series resonance frequency (f s,SH ) of the shunt resonators B SH  and above the parallel resonance frequency (f p,SER ) of the series resonators B SER . As noted previously, there is a constant need to improve the performance of acoustic resonators and filters that are based thereon. 
     SUMMARY 
     In one embodiment, a filter circuit includes a first input node and a second input node for receiving an input signal, and a first output node and a second output node for providing an output signal. A first series acoustic resonator is coupled in series between the first input node and the first output node. At least one coupled resonator filter (CRF) includes first and second transducers, which may be acoustically coupled to one another. The first transducer has a first electrode coupled to the first input node, a second electrode coupled to the second input node, and a first piezoelectric layer between the first electrode and the second electrode. A second transducer has a third electrode coupled to the first output node, a fourth electrode coupled to the second output node, and a second piezoelectric layer between the third electrode and the fourth electrode. The at least one CRF may have a first CRF that includes the first transducer, the second transducer, and a first coupling structure between the first transducer and the second transducer, wherein the first transducer and the second transducer are vertically aligned such that the first coupling structure vertically acoustically couples the first transducer and the second transducer. 
     Those skilled in the art will appreciate the scope of the present disclosure and realize additional aspects thereof after reading the following detailed description of the preferred embodiments in association with the accompanying drawing figures. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING FIGURES 
       The accompanying drawing figures incorporated in and forming a part of this specification illustrate several aspects of the disclosure, and together with the description serve to explain the principles of the disclosure. 
       The accompanying drawing figures incorporated in and forming a part of this specification illustrate several aspects of the disclosure and, together with the description, serve to explain the principles of the disclosure. 
         FIG. 1  illustrates a conventional Bulk Acoustic Wave (BAW) resonator. 
         FIG. 2  is a graph of the magnitude and phase of impedance over frequency responses as a function of frequency for an ideal BAW resonator. 
         FIGS. 3A-3C  are graphs of phase responses for various BAW resonator configurations. 
         FIG. 4  illustrates a conventional BAW resonator with a border ring. 
         FIG. 5A  is a schematic of a conventional ladder network. 
         FIGS. 5B and 5C  are graphs of a frequency response for BAW resonators in the conventional ladder network of  FIG. 5A  and a frequency response for the conventional ladder network of  FIG. 5A . 
         FIGS. 6A-6E  are circuit equivalents for the ladder network of  FIG. 5A  at the frequency points  1 ,  2 ,  3 ,  4 , and  5 , which are identified in  FIG. 5C . 
         FIG. 7  is a block diagram of a mobile terminal according to one embodiment. 
         FIG. 8  is a schematic of an RF front-end according to a first embodiment. 
         FIG. 9  illustrates a rigid PCB and a flexible PCB coupled together by multiple coaxial cables according to a first embodiment. 
         FIG. 10  is a schematic of an RF front-end according to a second embodiment. 
         FIG. 11  illustrates an acoustic resonator in parallel with a compensation circuit, which includes a single shunt acoustic resonator. 
         FIG. 12  is a graph that illustrates exemplary frequency responses for the acoustic resonator, compensation circuit, and overall circuit of  FIG. 11 . 
         FIG. 13  illustrates an acoustic resonator in parallel with a compensation circuit, which includes at least two shunt acoustic resonators, according to a first embodiment. 
         FIG. 14  is a graph that illustrates exemplary frequency responses for the acoustic resonator, compensation circuit, and overall circuit of  FIG. 13 . 
         FIG. 15  is a graph that compares actual frequency responses of the overall circuits of  FIGS. 11 and 13 . 
         FIG. 16  illustrates a plurality of parallel acoustic resonators in parallel with a compensation circuit, which includes at least two shunt acoustic resonators, according to a second embodiment. 
         FIG. 17  is a graph that illustrates first exemplary frequency responses for the acoustic resonator, compensation circuit, and overall circuit of  FIG. 16 . 
         FIG. 18  is a graph that illustrates second exemplary frequency responses for the acoustic resonator, compensation circuit, and overall circuit of  FIG. 16 . 
         FIGS. 19A through 19D  illustrate transformation of the T-circuit impedance architecture of the compensation circuit of  FIG. 13  to a π (pi) impedance model. 
         FIG. 20  illustrates the overall circuit of  FIG. 13  using the π (pi) impedance model of  FIG. 19D . 
         FIG. 21  is a graph illustrating the overall shunt impedance, Zres, according to one embodiment. 
         FIG. 22  is a graph illustrating the series equivalent impedance, ZA, according to one embodiment. 
         FIGS. 23A and 23B  are graphs over different frequency ranges illustrating the absolute or magnitude of series impedance ZS, the series equivalent impedance ZA, and overall series impedance ZAs, according to one embodiment. 
         FIG. 24  is a cross-section of a coupled resonator filter (CRF), according to one embodiment. 
         FIG. 25  is a simplified symbolic representation of the CRF of  FIG. 24 . 
         FIG. 26  is a first embodiment of an acoustically coupled filter. 
         FIG. 27  is a second embodiment of an acoustically coupled filter. 
         FIG. 28  is third embodiment of an acoustically coupled filter. 
         FIG. 29  is a fourth embodiment of an acoustically coupled filter. 
         FIG. 30  is a fifth embodiment of an acoustically coupled filter. 
         FIG. 31  is a sixth embodiment of an acoustically coupled filter. 
         FIG. 32  is a seventh embodiment of an acoustically coupled filter. 
         FIG. 33  is an eighth embodiment of an acoustically coupled filter. 
         FIG. 34  is a ninth embodiment of an acoustically coupled filter. 
         FIG. 35  is a tenth embodiment of an acoustically coupled filter. 
         FIG. 36  is an eleventh embodiment of an acoustically coupled filter. 
         FIG. 37  is a twelfth embodiment of an acoustically coupled filter. 
         FIG. 38  is a thirteenth embodiment of an acoustically coupled filter. 
         FIG. 39  is a fourteenth embodiment of an acoustically coupled filter. 
     
    
    
     DETAILED DESCRIPTION 
     The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the embodiments and illustrate the best mode of practicing the embodiments. Upon reading the following description in light of the accompanying drawing figures, those skilled in the art will understand the concepts of the disclosure and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims. 
     It will be understood that, although the terms first, second, etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first element could be termed a second element, and similarly, a second element could be termed a first element, without departing from the scope of the present disclosure. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items. 
     It will be understood that when an element such as a layer, region, or substrate is referred to as being “on” or extending “onto” another element, it can be directly on or extend directly onto the other element or intervening elements may also be present. In contrast, when an element is referred to as being “directly on” or extending “directly onto” another element, there are no intervening elements present. Likewise, it will be understood that when an element such as a layer, region, or substrate is referred to as being “over” or extending “over” another element, it can be directly over or extend directly over the other element or intervening elements may also be present. In contrast, when an element is referred to as being “directly over” or extending “directly over” another element, there are no intervening elements present. It will also be understood that when an element is referred to as being “connected” or “coupled” to another element, it can be directly connected or coupled to the other element or intervening elements may be present. In contrast, when an element is referred to as being “directly connected” or “directly coupled” to another element, there are no intervening elements present. 
     Relative terms such as “below” or “above” or “upper” or “lower” or “horizontal” or “vertical” may be used herein to describe a relationship of one element, layer, or region to another element, layer, or region as illustrated in the figures. It will be understood that these terms and those discussed previously are intended to encompass different orientations of the device in addition to the orientation depicted in the figures. 
     The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the disclosure. As used herein, the singular forms “a,” “an,” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises,” “comprising,” “includes,” and/or “including” when used herein specify the presence of stated features, integers, steps, operations, elements, and/or components but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. 
     Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. It will be further understood that terms used herein should be interpreted as having a meaning that is consistent with their meaning in the context of this specification and the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein. 
     As defined herein, the term “coupled” without being preceded with the adjective “acoustically” refers to an electrical coupling as opposed to an acoustic coupling. The term “acoustically coupled” refers to an acoustic coupling as opposed to an electrical coupling. Further, the phrase “about the same as” when referring to the series resonance frequency of two or more devices means that the series resonance frequencies of the devices are within 0.1% of each other. 
     While the concepts provided herein are applicable to various technologies, these concepts are particularly useful in mobile terminals, such as mobile telephones, tablets, computers, and like smart devices. The following provides an overview of such devices. Today&#39;s mobile terminals must communicate using different communication technologies in different bands, which vary significantly in both bandwidth and frequency. To further complicate matters, data rates are ever increasing and the there is a need to transmit and receive over these different bands at the same time. As a result, mobile terminals have very complicated front-end configurations and are starting to employ multiple input multiple output (MIMO) transmission and reception technology, which requires the use of multiple antennas. 
       FIG. 7  is a block diagram of a mobile terminal  42  that incorporates four antennas: a primary antenna A 1 , a secondary antenna A 2 , a tertiary antennae A 3 , and a quaternary antenna A 4 . The mobile terminal  42  generally includes control circuitry  44 , which is associated with a user interface (I/F)  46 , and radio frequency (RF) circuitry  48 . The user interface  46  may include microphones, speakers, keypads, touchscreens, displays, and the like. The RF circuitry  48  may include baseband, transceiver, power amplifier, and switching circuitry, as will be appreciated by those skilled in the art. 
     In general, signals to be transmitted are provided by the RF circuitry  48  to one or more of the antennas A 1  through A 4 , and signals received by one or more of the antennas A 1  through A 4  are routed to the RF circuitry  48  for demodulation and associated processing. The RF circuitry  48  may be configured to facilitate any number of communications, including first, second, third, fourth, and fifth generation cellular communications, wireless local area network (WLAN) communications, Bluetooth communications, industrial, scientific and medical (ISM) communications, near field communications, and the like. Any of these communications may use MIMO for transmission, reception, or both, depending on the capabilities of the mobile terminal  42  and the systems with which the mobile terminal  42  communicates. 
     Since mobile terminals  42  are relatively small, the multiple antennas A 1  through A 4  used for MIMO are relatively close to one another. As a result, the antennas A 1  through A 4  may interact with one another, and as a result, modify each other&#39;s radiation patterns, which generally alters the antenna&#39;s radiation efficiency. With continued reference to  FIG. 7 , when the primary antenna A 1  is used for transmission and the tertiary antennae A 3  is used for reception at the same time, the transmission from antennae A 1  may significantly degrade the ability to receive signals via antenna A 3 , given the proximity of antenna A 1  to antenna A 3 . Further, the secondary antenna A 2  and the quaternary antenna A 4  may also be impacted by transmissions from the primary antenna A 1 . As such, there is a need for a cost effective and space efficient technique to resolve, or at least significantly reduce, the impact that one antenna has on another in devices such as the mobile terminal  42  illustrated in  FIG. 7 . 
     With reference to  FIG. 8 , a technique for addressing the above issues is described. As illustrated, the RF circuitry  48  is associated with antennas A 1  through A 4 . Each of the antennas is associated with antenna tuning circuitry  50 ,  52 ,  54 ,  56 , respectively. In particular, antenna A 1  is coupled to the RF circuitry  48  through coaxial cable  58 . Antenna A 3  is coupled to the RF circuitry  48  through coaxial cable  60  and a MPMS filter  62 ; antenna A 2  is coupled to the RF circuitry  48  through coaxial cable  64  and MPMS filter  66 ; and antenna A 4  is coupled to the RF circuitry  48  through coaxial cable  68  and MPMS filter  70 . While MPMS filters  62 ,  66 , and  70  are provided for antennas A 3 , A 2 , and A 4 , respectively, alternative embodiments may only employ MPMS filter  62 , given the proximity of antennas A 1  and A 3 . In other embodiments, an appropriately configured MPMS filter (not illustrated) may also be provided in association with antenna A 1 . In short, MPMS filters may be provided for each antenna A 1  through A 4  or any combination thereof. The antenna tuning circuitry  50 ,  52 ,  54 ,  56  are used for tuning impedances associated with the respective antennas A 1 , A 3 , A 2 , A 4 , as those skilled in the art will appreciate. 
     For the following description, MPMS filter  62  is described in detail; however, MPMS filter  66  and  70  may be similarly or identically configured, depending on the embodiment. Assume that the RF circuitry  48  is configured to transmit RF signals in band X and band Y via antenna A 1  at the same or different times. Further assume that RF circuitry  48  is configured to receive RF signals and bands A, B, and C at the same or different times. Given the proximity of antennas A 1  and A 3 , transmitting in bands X or Y via antenna A 1  would significantly impact the ability of antenna A 3  to receive RF signals in bands A, B, or C, in the absence of MPMS filter  62 . However, adding MPMS filter  62  in close proximity to antenna A 3  significantly reduces the impact that antenna A 1  has on antenna A 3 . 
     MPMS filter  62  is a specially configured filter that has multiple passbands and multiple stopbands, which are interleaved with one another, as illustrated in  FIG. 8 . In this example, passbands are provided for bands A, B, and C and stopbands are provided for at least bands X and Y. A stopband for band X is between passbands for bands A and B, and a stopband for band Y is between passbands for bands B and C. In other words, stopbands are provided for the problematic bands that are transmitted via antenna A 1 , and passbands are provided for the bands to be received via antenna A 3 . The RF circuitry  48  may also transmit signals in bands A, B, or C via antenna A 3 . Providing stopbands for an adjacent antenna&#39;s transmission bands and passbands for the selected antenna&#39;s receive (and transmission) bands can significantly improve the performance of both antennas. When multiple ones of the MPMS filters  62 ,  66 ,  70  are employed, the passbands and stopbands may be the same or different amongst the different MPMS filters  62 ,  66 ,  70 , based on the proximity of the antennas A 1 -A 4  as well as the communication bands used for communications by the mobile terminal  42 . 
     In certain embodiments, at least two of the stopbands and/or passbands provided by one or more of the MPMS filters  62 ,  66 ,  70  reside entirely above 2 GHz and have a bandwidth of at least 20 MHz. In other embodiments, at least two of the stopbands and/or passbands reside between 2 GHz and 12 GHz and have a bandwidth of at least 20 MHz, 40 MHz, 50 MHz, or 100 MHz. In select embodiments, at least one of the stopbands or passbands residing between two other stopbands or passbands has a bandwidth of at least 100 MHz, 150 MHz, or 200 MHz. All, or at least certain of, the stopbands may provide attenuation of at least 10 dB, 20 dB, or 30 dB in each of the above embodiments, depending on the configuration of the MPMS filters  62 ,  66 ,  70 . 
     With reference to  FIG. 9 , the mobile terminal  42  may employ multiple printed circuit boards (PCBs) to implement the necessary electronics for operation. Further, the various antennas A 1 -A 4  may be spread about the mobile terminal  42 . These antennas A 1 -A 4  may be implemented on or in a housing H (illustrated in  FIG. 7 ) of the mobile terminal  42 , on the various PCBs, or a combination thereof.  FIG. 9  illustrates a rigid PCB (R-PCB) and a flexible PCB (F-PCB), which are used to implement at least part of the electronics for the mobile terminal  42 . In one embodiment, the rigid PCB R-PCB may be a traditional glass-reinforced multilayer circuit board, wherein the flexible PCB F-PCB is provided by a much thinner, flexible substrate on which traces and components may be formed or mounted. The flexible PCB F-PCB will have a flex factor of at least ten times that of the rigid PCB R-PCB. 
     As illustrated, the control circuitry  44  and the RF circuitry  48  are implemented in whole or in part on the rigid PCB R-PCB while the MPMS filters  62 ,  66 ,  70  and the antenna tuning circuitry  50 ,  52 ,  54 ,  56  are implemented on the flexible PCB F-PCB. The coaxial cables  58 ,  60 ,  64 ,  68  connect the rigid PCB R-PCB and the flexible PCB F-PCB such that the transmit/receive paths that extend between the RF circuitry  48  and the respective antennas A 1 , A 2 , A 3 , and A 4  are provided by the combination of the rigid PCB R-PCB, the flexible PCB F-PCB, and the coaxial cables  58 ,  60 ,  64 ,  68 . These transmit/receive paths extend to corresponding antenna ports AP 1 , AP 2 , AP 3 , AP 4  of the flexible PCB F-PCB. The antennas A 1 , A 2 , A 3 , and A 4  are connected to the antenna ports AP 1 , AP 2 , AP 3 , AP 4 , respectively, through cables, traces, and/or the like. 
     With reference to  FIG. 10 , a low noise amplifier (LNA)  72  may be provided between the MPMS filter  62  and the coaxial cable  60  to amplify the filtered receive signals prior to the coaxial cable  60 . The LNA  72  may be provided along with the MPMS filter  62  on the flexible PCB F-PCB, wherein the RF circuitry  48  is provided on the rigid PCB R-PCB, and the coaxial cable  60  connects the flexible PCB F-PCB and the rigid PCB R-PCB. The antenna tuning circuitry  50 ,  52  may also be provided on the flexible PCB F-PCB. 
     The following provides various filters that employ acoustic resonators and are capable of providing a filter response that includes multiple passbands and stopbands. Some basics regarding the theory of operation are provided prior to describing the specific configurations, which provide the desired filter responses. 
     Turning now to  FIG. 11 , a series resonator B 1  is shown coupled between an input node I/P and an output node O/P. The series resonator B 1  has a series resonance frequency F s  and inherent capacitance, which generally limits the bandwidth of filters that employ the series resonator B 1 . In the case of a Bulk Acoustic Wave (BAW) resonator, the capacitance of the series resonator B 1  is primarily caused by its inherent structure, which looks and acts like a capacitor in part because the series resonator includes the top and bottom electrodes  20 ,  22  ( FIG. 1 ) that are separated by a dielectric piezoelectric layer  18 . While BAW resonators are the focus of the example, other types of acoustic resonators, such as Surface Acoustic Wave (SAW) resonators, are equally applicable. 
     A compensation circuit  74  is coupled in parallel with the series resonator B 1  and functions to compensate for some of the capacitance presented by the series resonator B 1 . The compensation circuit  74  includes two negatively coupled inductors L 1 , L 2  and a shunt resonator B 2 . The inductors L 1 , L 2  are coupled in series between the input node I/P and the output node O/P, wherein a common node CN is provided between the inductors L 1 , L 2 . The inductors L 1 , L 2  are magnetically coupled by a coupling factor K, wherein the dots illustrated in association with the inductors L 1 , L 2  indicate that the magnetic coupling is negative. As such, the inductors L 1 , L 2  are connected in electrical series and negatively coupled from a magnetic coupling perspective. As defined herein, two (or more) series-connected inductors that are negatively coupled from a magnetic perspective are inductors that are:
         connected in electrical series; and   the mutual inductance between the two inductors functions to decrease the total inductance of the two (or more) inductors.
 
The shunt resonator B 2  is coupled between the common node CN and ground, or other fixed voltage node.
       

     To compensate for at least some of the capacitance of the series resonator B 1 , the compensation circuit  74  presents itself as a negative capacitance within certain frequency ranges when coupled in parallel with the series resonator B 1 . Since capacitances in parallel are additive, providing a negative capacitance in parallel with the (positive) capacitance of the series resonator B 1  effectively reduces the capacitance of the series resonator B 1 . With the compensation circuit  74 , the series resonator B 1  can actually function as a filter (instead of just a resonator) and provide a passband, albeit a fairly narrow passband, instead of a more traditional resonator response (solid line of  FIG. 2 ). 
       FIG. 12  graphically illustrates the frequency responses of the series resonator B 1  (inside the block referenced B 1 ), the compensation circuit  74  (inside the block referenced  74 ), and the overall circuit in which the compensation circuit  74  is placed in parallel with the series resonator B 1 . As illustrated, the overall circuit provides a relatively narrow passband. Further detail on this particular circuit topology can be found in the co-assigned U.S. patent application Ser. No. 15/004,084, filed Jan. 22, 2016, now patented as U.S. Pat. No. 9,837,984 on Dec. 5, 2017, and titled RF LADDER FILTER WITH SIMPLIFIED ACOUSTIC RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, and U.S. patent application Ser. No. 14/757,651, filed Dec. 23, 2015, now patented as U.S. Pat. No. 10,333,494 on Jun. 25, 2019, and titled SIMPLIFIED ACOUSTIC RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, the disclosures of which are incorporated herein by reference in their entireties. 
     While beneficial in many applications, the narrow passband of the circuit topology of  FIG. 11  has its limitations. With the challenges of modern day communication systems, wider passbands and the ability to provide multiple passbands within a given system are needed. Fortunately, applicants have discovered that certain modifications to this topology provide significant and truly unexpected increases in passband bandwidths and, in certain instances, the ability to generate multiple passbands of the same or varying bandwidths in an efficient and effective manner. 
     With reference to  FIG. 13 , a modified circuit topology is illustrated wherein the circuit topology of  FIG. 11  is modified to include an additional shunt resonator B 3 , which is coupled between the common node CN and ground. As such, a new compensation circuit  76  is created that includes the negatively coupled inductors L 1  and L 2 , which have a coupling coefficient K, and at least two shunt resonators B 2 , B 3 . The compensation circuit  76  is coupled in parallel with the series resonator B 1 . When the series resonance frequencies F s  of the shunt resonators B 2 , B 3  are different from one another, unexpectedly wide bandwidths passbands are achievable while maintaining a very flat passbands, steep skirts, and excellent cancellation of signals outside of the passbands. 
       FIG. 14  graphically illustrates the frequency responses of the series resonator B 1  (inside the block referenced B 1 ), the compensation circuit  76  (inside the block referenced  76 ), and the overall circuit in which the compensation circuit  76  is placed in parallel with the series resonator B 1 . As illustrated, the overall circuit with the compensation circuit  76  provides a much wider passband ( FIG. 10 ) than the overall circuitry with the compensation circuit  74  ( FIG. 12 ). 
     While  FIGS. 12 and 14  are graphical representations,  FIG. 15  is an actual comparison of the frequency response of the overall circuit using the different compensation circuits  74 ,  76 , wherein the overall circuit using the compensation circuit  76  provides a significantly wider and better formed passband (solid line) than the overall circuit using the compensation circuit  74  (dashed line). 
     As illustrated in  FIG. 16 , the concepts described herein not only contemplate the use of multiple shunt resonators B 2 , B 3 , which are coupled between the common node CN and ground, but also multiple series resonators, such as series resonators B 1  and B 4 , which are coupled in parallel with one another between the input node VP and the output node O/P. The series resonance frequencies F s  of the series resonators B 1 , B 4  are different from one another, and the series resonance frequencies F s  of the shunt resonators B 2 , B 3  are also different from one another and different from those of the series resonators. While only two series resonators B 1 , B 4  and two shunt resonators B 2 , B 3  are illustrated, any number of these resonators may be employed depending on the application and the desired characteristics of the overall frequency response of the circuit in which these resonators and associated compensation circuits  76  are employed. While the theory of operation is described further below,  FIGS. 17 and 18  illustrate just two of the many possibilities. 
     For  FIG. 17 , there are two series resonators B 1 , B 4  and two shunt resonators B 2 , B 3 , with different and relatively dispersed series resonance frequencies F s .  FIG. 17  graphically illustrates the frequency responses of the combination of the two series resonators B 1 , B 4  (inside the block referenced BX), the compensation circuit  76  with two shunt resonators B 2 , B 3  (inside the block referenced  76 ), and the overall circuit in which the compensation circuit  76  is placed in parallel with the series resonator B 1 . As illustrated, the overall circuit in this configuration has the potential to provide a passband that is even wider than that for the embodiment of  FIGS. 13 and 14 . For example, passbands of greater than 100 MHz, 150 MHz, 175 MHz, and 200 MHz are contemplated at frequencies at or above 1.5 GHz, 1.75 GHz, and 2 GHz. 
     In other words, center-frequency-to-bandwidth ratios (fc/BW*100) of 3.5% to 9%, 12%, or greater are possible, wherein fc is the center frequency of the passband and BW is the bandwidth of the passband. If multiple passbands are provided, BW may encompass all of the provided passbands. Further, when multiple passbands are provided, the passbands may have the same or different bandwidths or center-frequency-to-bandwidth ratios. For example, one passband may have a relatively large center-frequency-to-bandwidth ratio, such as 12%, and a second passband may have a relatively small center-frequency-to-bandwidth ratio, such as 2%. Alternatively, multiple ones of the passbands may have a bandwidth of 100 MHz, or multiple ones of the passbands may have generally the same center-frequency-to-bandwidth ratios. In the latter case, the bandwidths of the passbands may inherently be different from one another, even though the center-frequency-to-bandwidth ratios are the same. 
     For  FIG. 18 , there are four series resonators, which are coupled in parallel with one another (not shown), and two shunt resonators (not shown) with different and more widely dispersed series resonance frequencies F s .  FIG. 18  graphically illustrates the frequency responses of the combination of the four series resonators (inside the block referenced BX), the compensation circuit  76  with two shunt resonators B 2 , B 3  (inside the block referenced  76 ), and the overall circuit in which the compensation circuit  76  is placed in parallel with the series resonator B 1 . As illustrated, the overall circuit in this configuration provides multiple passbands, which are separated by a stopband. In this embodiment, two passbands are provided; however, the number of passbands may exceed two. The number of passbands in the bandwidth of each of the passbands is a function of the number of shunt and series resonators B 1 -B 4  and the series resonance frequencies F s  thereof. 
     The theory of the compensation circuit  76  follows and is described in association with  FIGS. 19A through 19D  and  FIG. 20 . With reference to  FIG. 19A , assume the compensation circuit  76  includes the two negatively coupled inductors L 1 , L 2 , which have an inductance value L, and two or more shunt resonators BY, which have an overall shunt impedance Zres presented between the common node CN and ground. While the inductance values L of the negatively coupled inductors L 1 , L 2  are described as being the same, these values may differ depending on the application. Also assume that the one or more series resonators BX present an overall series impedance ZS. 
     As shown in  FIG. 19B , the two negatively coupled and series-connected inductors L 1 , L 2  (without Zres) can be modeled as a T-network of three inductors L 3 , L 4 , and L 5 , wherein series inductors L 3  and L 4  are connected in series and have a value of L(1+K), and shunt inductor L 5  has a value of −L*K, where K is a coupling factor between the negatively coupled inductors L 1 , L 2 . Notably, the coupling factor K is a positive number between 0 and 1. Based on this model, the overall impedance of the compensation circuit  76  is modeled as illustrated in  FIG. 19C , wherein the shunt impedance Zres is coupled between the shunt inductor L 5  and ground. The resulting T-network, as illustrated in  FIG. 19C , can be transformed into an equivalent π (pi) network, as illustrated in  FIG. 19D . 
     The π network of  FIG. 19D  can be broken into a series impedance ZA and two shunt equivalent impedances ZB. The series equivalent impedance ZA is represented by two series inductances of value L*(1+K), where K&gt;0, and a special “inversion” impedance Zinv. The inversion impedance Zinv is equal to [L(1+K)ω] 2 /[Zres−jLKω], where ω=2πf and f is the frequency. As such, the series equivalent impedance ZA equals j*2*L(1+K)ω+Zinv and is coupled between the input node I/O and the output node O/P. Each of the two shunt equivalent impedances ZB is represented by an inductor of value L(1−K) in series with two overall shunt impedances Zres. 
     Notably, the series equivalent impedance ZA has a negative capacitor behavior at certain frequencies at which broadband cancellation is desired and has series resonance at multiple frequencies. In general, the series equivalent impedance ZA has a multiple bandpass-bandstop characteristic, in that the series equivalent impedance ZA will pass some frequencies and stop others. When the series equivalent impedance ZA is placed in parallel with the series impedance ZS of the series resonators BX, which can also have a multiple bandpass-bandstop characteristic, a broadband filter or a filter with multiple passbands may be created. 
       FIG. 20  illustrates the series impedance ZS of the series resonators BX in parallel with the series equivalent impedance ZA of the compensation circuit  76 . The overall series impedance ZAs represents the series impedance ZS in parallel with the series equivalent impedance ZA. The two shunt impedances ZB are respectively coupled between the input port I/P and ground and the output port O/P and ground. The primary focus for the following discussion relates to the series equivalent impedance ZA and its impact on the series impedance ZS when the series equivalent impedance ZA is placed in parallel with the series impedance ZS. 
     As noted previously, the series equivalent impedance ZA provides two primary functions. The first provides a negative capacitive behavior, and the second provides one or more additional series resonances between the input node I/P and the output node O/P. These additional series resonances are provided through the series equivalent impedance ZA and are in addition to any series resonances that are provided through the series impedance ZS of the series resonators BX. To help explain the benefits and concept of the negative capacitive behavior provided by the series equivalent impedance ZA, normal capacitive behavior is illustrated in association with the overall shunt impedance Zres, which is provided by the shunt resonators BY.  FIG. 21  graphs the absolute (magnitude) and imaginary components of the overall shunt impedance Zres, which is formed by two shunt resonators BY coupled in parallel with one another. 
     The series resonance frequency F s  for each of the two shunt resonators BY occurs when the absolute impedance (abs(Zres)) is at or near zero. Since there are two shunt resonators BY, the absolute impedance (abs(Zres)) is at or near zero at two frequencies, and as such, there are two series resonance frequencies F s . The parallel resonance frequencies F p  occur when the imaginary component (imag(Zres)) peaks. Again, since there are two shunt resonators BY, there are two series resonance frequencies F s  provided by the overall shunt impedance Zres. 
     Whenever the imaginary component (imag(Zres)) of the overall shunt impedance Zres is less than zero, the overall shunt impedance Zres has a capacitive behavior. The capacitive behavior is characterized in that the reactance of the overall shunt impedance Zres is negative and decreases as frequency increases, which is consistent with capacitive reactance, which is represented by 1/jωC. The graph of  FIG. 21  identifies three regions within the impedance response of the overall shunt impedance Zres that exhibit capacitive behavior. 
     Turning now to  FIG. 22 , the series equivalent impedance ZA is illustrated over the same frequency range as that of the overall shunt impedance Zres, illustrated in  FIG. 21 . The series equivalent impedance ZA has two series resonance frequencies F s , which occur when the absolute impedance (abs(ZA)) is at or near zero. The two series resonance frequencies F s  for the series equivalent impedance ZA are different from each other and slightly different from those for the overall shunt impedance Zres. Further, the number of series resonance frequencies F s  generally corresponds to the number of shunt resonators BY in the compensation circuit  76 , assuming the series resonance frequencies F s  are different from one another. 
     Interestingly, the imaginary component (imag(ZA)) of the series equivalent impedance ZA is somewhat inverted with respect to that of the overall shunt impedance Zres. Further, the imaginary component (imag(ZA)) of the series equivalent impedance ZA has a predominantly positive reactance. During the portions at which the imaginary component (imag(ZA)) is positive, the reactance of the series equivalent impedance ZA again decreases as frequency increases, which is indicative of capacitive behavior. However, the reactance is positive, whereas traditional capacitive behavior would present a negative reactance. This phenomenon is referred to as negative capacitive behavior. Those portions of the imaginary component (imag(ZA)) of the series equivalent impedance ZA that are positive and thus exhibit negative capacitive behavior are highlighted in the graph of  FIG. 22 . 
     The negative capacitive behavior of the series equivalent impedance ZA for the compensation circuit  76  is important, because when the series equivalent impedance ZA is placed in parallel with the series impedance ZS, the effective capacitance of the overall circuit is reduced. Reducing the effective capacitance of the overall circuit shifts the parallel resonance frequency F p  of the series impedance ZS higher in the frequency range, which is described subsequently, and significantly increases the available bandwidth for passbands while providing excellent out-of-band rejection. 
     An example of the benefit is illustrated in  FIGS. 23A and 23B . The thicker solid line, which is labeled abs(VG), represents the frequency response of the overall circuit illustrated in  FIG. 16 , wherein there are two series resonators BX and two shunt resonators BY in the compensation circuit  76 . The frequency response has two well-defined passbands, which are separated by a stopband. The frequency response abs(VG) of the overall circuit generally corresponds to the inverse of the overall series impedance ZAs, which again represents the series impedance Zs in parallel with the series equivalent impedance ZA, as provided in  FIG. 20 . 
     Notably, the parallel resonance frequencies F p (ZS) of the series impedance ZS, in isolation, fall in the middle of the passbands of frequency response abs(VG) of the overall circuit. If the parallel resonance frequencies F p (ZS) of the series impedance ZS remained at these locations, the passbands would be severely affected. However, the negative capacitive behavior of the series equivalent impedance ZA functions to shift these parallel resonance frequencies F p (ZS) of the series impedance ZS to a higher frequency and, in this instance, above the respective passbands. This is manifested in the resulting overall series impedance ZAs, in which the only parallel resonance frequencies F p (ZAs) occur above and outside of the respective passbands. An additional benefit to having the parallel resonance frequencies F p (ZAs) occur outside of the respective passbands is the additional cancellation of frequencies outside of the passbands. Plus, the overall series impedance ZAs is lower than the series impedance ZS within the respective passbands. 
     A further contributor to the exemplary frequency response abs(VG) of the overall circuit is the presence of the additional series resonance frequencies F s , which are provided through the series equivalent impedance ZA. These series resonance frequencies F s  are offset from each other and from those provided through the series impedance ZS. The series resonance frequencies F s  for the series equivalent impedance ZA in the series impedance ZS occur when the magnitudes of the respective impedances approach zero. The practical results are wider passbands, steeper skirts for the passbands, and greater rejection outside of the passbands, as evidenced by the frequency response abs(VG) of the overall circuit. 
     Turning now to  FIG. 24 , a cross-section of a coupled resonator filter (CRF)  80  is illustrated. The CRF  80  is essentially a BAW device including two or more vertically stacked transducers. As with the BAW resonator  10  of  FIG. 1 , the CRF  80  has a substrate  82 , a reflector  84 , which includes multiple reflector layers  84 L, and a bottom transducer  86 . The bottom transducer  86  includes a top electrode  88 , a bottom electrode  90 , and a bottom piezoelectric layer  92  sandwiched therebetween. Unlike the BAW resonator  10 , a coupling structure  94  is provided over the top electrode  88  of the bottom transducer  86 . The coupling structure  94  includes multiple coupling layers  96 , which are typically layers of alternating low and high acoustic impedances. A top transducer  98  is provided over the coupling structure  94  and includes a top electrode  100 , a bottom electrode  102 , and a top piezoelectric layer  104 , which is sandwiched between the top electrode  100  and the bottom electrode  102 . 
     The coupling structure  94  functions to acoustically couple the top transducer  98  and the bottom transducer  86  for one or more acoustic wavelengths or ranges thereof. While the materials may vary, a coupling structure  94  that includes three coupling layers  96  could include alternating layers of oxide, tungsten, and oxide, respectively. The coupling layers  96  may have thicknesses corresponding to one quarter of the acoustic wavelength for the frequency or frequencies of coupling. A reduced complexity block representation of the CRF  80  is provided in  FIG. 25 , and is featured prominently in the embodiments discussed below. For further information regarding the functionality and structure of the CRF  80 , reference is made to Lakin, K. M. (2002). Coupled Resonator Filters. Proceedings of the IEEE Ultrasonics Symposium. 1. 901-908 vol. 1. 10.1109/ULTSYM.2002.1193543; Shirakawa, Alexandre &amp; Thalhammer, Robert &amp; Jamneala, T &amp; B. Koelle, Uli. (2011). Bulk Acoustic Wave-Coupled Resonator Filters: Concept, Design, and Application. International Journal of RF and Microwave Computer-Aided Engineering. 21. 477-485. 10.1002/mmce.20552; and U.S. Pat. No. 6,720,844, which are incorporated herein by reference in their entireties. 
     For the embodiments described below, the compensation circuit  76  that includes the inductors L 1  and L 2  is replaced with one or more CRFs  80  to achieve the same or similar functionality.  FIG. 26  illustrates a filter circuit where an input signal I/P is provided at terminals S 1  and S 2  and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. The top electrode  100  of the top transducer  98  is coupled to terminal S 1 , and the bottom electrode  102  of the top transducer  98  is coupled to terminal S 2 . The top electrode  88  of the bottom transducer  86  is coupled to terminal S 4 , and the bottom electrode  90  of the bottom transducer  86  is coupled to terminal S 3 . 
     As such, the input signal drives the top transducer  98 , which is acoustically coupled to the bottom transducer  86 . The acoustic coupling will result in an electrical signal being generated between the top electrode  88  and the bottom electrode  90  of the bottom transducer  86 . The portion or portions of the input signal that are coupled from the top transducer  98  to the bottom transducer  86  are effectively inverted (180 degrees phase-shifted) and presented to terminals S 3 , S 4  for combining with those portions of the input signal that are passed through the BAW resonators B 5 , B 6 . As a result, the filter circuit illustrated in  FIG. 26 , which includes the CRF  80 , is capable of functioning in a similar fashion to that illustrated in  FIG. 16 , which includes the compensation circuit  78 . The inversion for the acoustically coupled embodiments is provided by effectively inverting the output taken off of one of the transducers  86 ,  98  relative to the input. 
     Notably, the circuit of  FIG. 24  and those that follow have a similar topology and may provide similar functionality to a ladder network, such as that illustrated in  FIG. 5A . The top transducer  98  corresponds to a shunt transducer or resonator that extends between the input terminals S 1  and S 2 , the bottom transducer  86  corresponds to another shunt transducer or resonator that extends between the output terminals S 3  and S 4 , and the BAW resonators B 5 , B 6  are series acoustic resonators that extend between input terminal S 1  and output terminal S 3 . The key difference is that the acoustic coupling between the top and bottom transduces  98 ,  86  provides designers additional parameters for fine tuning and improving the performance of such filter circuits in a cost and space effective manner. Connections of the various electrodes  88 ,  90 ,  100 ,  102  to the terminals S 1 , S 2 , S 3 , S 4  may be such that the coupled frequencies are coupled in or out of phase depending on whether the supplemental acoustic energies are intended to combine with one another or cancel one another. The concepts herein provide tremendous flexibility for new design techniques. 
     Depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. Further, the series resonance frequencies of the top and bottom transducers  98 ,  86  may, and will likely, be different from the series resonance frequencies of BAW resonators B 5 , B 6 . Further, the series resonance frequencies of the BAW resonators B 5 , B 6  may, and likely will, be different from each other. 
     Turning now to  FIG. 27 , the filter circuit includes a first CRF  80  and a second CRF  106 . An input signal I/P is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrode  100  of the top transducer  98  of CRF  80  is coupled to terminal S 1  as well as to the top electrode  100  of the top transducer  98  of CRF  106 . The bottom electrode  102  of the top transducer  98  of CRF  80  is coupled to terminal S 2  as well as to the bottom electrode  102  of the top transducer  98  of CRF  106 . The top electrode  88  of the bottom transducer  86  of CRF  106  is coupled to terminal S 4  as well as to the top electrode  88  of the bottom transducer  86  of CRF  80 . The bottom electrode  90  of the bottom transducer  86  of CRF  106  is coupled to terminal S 3  as well as to the bottom electrode  90  of the bottom transducer  86  of CRF  80 . 
     For each CRF  80 ,  106 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may, and will likely, be different between the CRFs  80 ,  106 . For example, the top and bottom transducers  98 ,  86  for CRF  80  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  106  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . Alternatively, the top transducer  98  for CRF  80  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  80  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  106  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  106  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
     In  FIG. 28 , the parallel BAW resonators B 5 , B 6  are replaced with a CRF  108 . Again, an input signal I/P is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . Terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. Terminal S 1  is coupled to the top electrode  100  of the top transducer  98  of CRF  108  as well as to the top electrode  100  of the top transducer  98  of CRF  80 . Terminal S 2  is coupled to the bottom electrode  102  of the top transducer  98  of CRF  108  as well as to the bottom electrode  102  of the top transducer  98  of CRF  80 . 
     Terminal S 3  is coupled to the top electrode  88  of the bottom transducer  86  of CRF  108  as well as to the bottom electrode  90  of the bottom transducer  86  of CRF  80 . Terminal S 4  is coupled to the bottom electrode  90  of the bottom transducer  86  of CRF  108  as well as to the top electrode  88  of the bottom transducer  86  of CRF  80 . As such, the signal generated across the bottom transducer  86  of CRF  80  is inverted relative to the signal generated across the bottom transducer  86  of CRF  108 . 
     For each CRF  80 ,  108 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may, and will likely, be different between the CRFs  80 ,  108 . For example, the top and bottom transducers  98 ,  86  for CRF  80  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  108  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . Alternatively, the top transducer  98  for CRF  80  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  80  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  108  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  108  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
     Another embodiment is illustrated in  FIG. 29 . In this embodiment, four CRFs  80 ,  106 ,  110 ,  112  are provided. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . Terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. Terminal S 1  is coupled to the top electrodes  100  of the top transducers  98  of CRFs  80 ,  106 ,  110 ,  112 . Terminal S 2  is coupled to the bottom electrodes  102  of the top transducers  98  of CRFs  80 ,  106 ,  110 ,  112 . 
     Terminal S 3  is coupled to the top electrodes  88  of the bottom transducers  86  of CRFs  110 ,  112  as well as to the bottom electrodes  90  of the bottom transducers  86  of CRFs  80 ,  106 . Terminal S 4  is coupled to the bottom electrodes  90  of the bottom transducers  86  of CRF  110 ,  112  as well as to the top electrodes  88  of the bottom transducers  86  of CRFs  80 ,  106 . As such, the signals generated across the bottom transducers  86  of CRFs  80  and  106  are inverted relative to the signals generated across the bottom transducers  86  of CRFs  110 ,  112 . 
     For each CRF  80 ,  106 , 110 ,  112  depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may, and will likely, be different between the CRFs  80 ,  106 ,  110 ,  112 . For example, the top and bottom transducers  98 ,  86  for CRF  80  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  106  may have a series resonance frequency of FS 2 , the top and bottom transducers  98 ,  86  for CRF  110  may have a series resonance frequency of FS 3 , and the top and bottom transducers  98 ,  86  for CRF  112  may have a series resonance frequency of FS 4 , wherein F 1 , F 2 , F 3 , and F 4  are unique series resonance frequencies. Alternatively, the top transducer  98  for one or more of the CRFs  80 ,  106 ,  110 ,  112  may have a different series resonance frequency. 
     A variant of the filter circuit of  FIG. 28  is provided  FIG. 30 . In this embodiment, an input signal I/P is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . Terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. Terminal S 1  is coupled to the top electrode  100  of the top transducer  98  of CRF  108  as well as to the top electrode  88  of the bottom transducer  86  of CRF  80 . Terminal S 2  is coupled to the bottom electrode  102  of the top transducer  98  of CRF  108  as well as to the bottom electrode  90  of the bottom transducer  86  of CRF  80 . 
     Terminal S 3  is coupled to the top electrode  88  of the bottom transducer  86  of CRF  108  as well as to the bottom electrode  102  of the top transducer  98  of CRF  80 . Terminal S 4  is coupled to the bottom electrode  90  of the bottom transducer  86  of CRF  108  as well as to the bottom electrode  102  of the top transducer  98  of CRF  80 . As such, the signal generated across the bottom transducer  86  of CRF  80  is inverted relative to the signal generated across the top transducer  98  of CRF  108 . 
     For each CRF  80 ,  108 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may, and will likely, be different between the CRFs  80 ,  108 . For example, the top and bottom transducers  98 ,  86  for CRF  80  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  108  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . Alternatively, the top transducer  98  for CRF  80  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  80  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  108  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  108  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
     A variant of the filter circuit of  FIG. 29  is provided  FIG. 31 . In this embodiment, four CRFs  80 ,  106 ,  110 ,  112  are provided. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . Terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. Terminal S 1  is coupled to the top electrodes  100  of the top transducers  98  of CRFs  110 ,  112  as well as to the top electrodes  88  of the bottom transducers  86  of CRFs  80 ,  106 . Terminal S 2  is coupled to the bottom electrodes  102  of the top transducers  98  of CRFs  110 ,  112  as well as to the bottom electrodes  90  of CRFs  80 ,  106 . 
     Terminal S 3  is coupled to the top electrodes  88  of the bottom transducers  86  of CRFs  110 ,  112  as well as to the bottom electrodes  102  of the top transducers  98  of CRFs  80 ,  106 . Terminal S 4  is coupled to the bottom electrodes  90  of the bottom transducers  86  of CRF  110 ,  112  as well as to the top electrodes  100  of the top transducers  98  of CRFs  80 ,  106 . As such, the signals generated across the bottom transducers  86  of CRFs  80  and  106  are inverted relative to the signals generated across the top transducers  98  of CRFs  110 ,  112 . 
     For each CRF  80 ,  106 ,  110 ,  112  depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may, and will likely, be different between the CRFs  80 ,  106 ,  110 ,  112 . For example, the top and bottom transducers  98 ,  86  for CRF  80  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  106  may have a series resonance frequency of FS 2 , the top and bottom transducers  98 ,  86  for CRF  110  may have a series resonance frequency of FS 3 , and the top and bottom transducers  98 ,  86  for CRF  112  may have a series resonance frequency of FS 4 , where F 1 , F 2 , F 3 , and F 4  are unique series resonance frequencies. 
     In another example, the top transducers  98  for CRFs  80 ,  110  may have a series resonance frequency of FS 1 , the bottom transducers  86  for CRFs  80 ,  110  may have a series resonance frequency of FS 2 , the top transducers  98  for CRFs  106 ,  112  may have a series resonance frequency of FS 3 , and the bottom transducers  86  for CRFs  106 ,  112  may have a series resonance frequency of FS 4 , where F 1 , F 2 , F 3 , and F 4  are unique series resonance frequencies. Other combinations are envisioned. 
     In the embodiments described above, vertical acoustic coupling occurs between the top transducer  98  and the bottom transducer  86  of the various CRFs  80 ,  106 ,  108 ,  110 ,  112 . Any lateral, or transversal coupling between the transducers  86 ,  98  of adjacent CRFs  80 ,  106 ,  108 ,  110 ,  112  is minimal, if nonexistent. In the following embodiments, transversal, or lateral coupling occurs between certain transducers of different CRFs  80  or BAW resonators  10 . 
     With reference to  FIG. 32 , a basic embodiment that employs transversal acoustic coupling is illustrated. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     Two transducers  114 ,  116  are transversally acoustically coupled by sharing a piezoelectric layer  118  (shown) or other coupling material (not shown). Transducer  114  includes a portion (leftmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  120  and a bottom electrode  122 . 
     Transducer  116  includes a portion (rightmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  124  and a bottom electrode  126 . 
     Terminal S 1  is coupled to the top electrode  120  of transducer  114 . Terminal S 2  is coupled to the bottom electrode  122  of transducer  114 . Terminal S 3  is coupled to the bottom electrode  126  of transducer  116 . Terminal S 4  is coupled to the top electrode  124  of transducer  116 . As such, the output of transducer  116  is inverted relative to the input of transducer  114 . The series resonance frequencies of the respective transducers  114 ,  116  may be the same or different depending on the application. 
     The embodiment of  FIG. 33  builds on that of  FIG. 32 . An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     A first pair of transducers  114 ,  116  are transversally acoustically coupled by sharing a piezoelectric layer  118  and are coupled between the input terminals S 1 , S 2  and the output terminals S 3 , S 4 . A second pair of transducers  128 ,  130  are transversally acoustically coupled by sharing a piezoelectric layer  118  and are coupled between the input terminals S 1 , S 2  and the output terminals S 3 , S 4 . The first pair of transducers  114 ,  116  is effectively coupled in parallel with the second pair of transducers  128 ,  130 . 
     Transducer  114  includes a portion (leftmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  120  and a bottom electrode  122 . Transducer  116  includes a portion (rightmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  124  and a bottom electrode  126 . Similarly, transducer  128  includes a portion (leftmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  120  and a bottom electrode  122 . Transducer  130  includes a portion (rightmost portion) of the piezoelectric layer  118  sandwiched between a top electrode  124  and a bottom electrode  126 . 
     Terminal S 1  is coupled to the top electrodes  120  of transducers  114 ,  128 . Terminal S 2  is coupled to the bottom electrodes  122  of transducers  114 ,  128 . Terminal S 3  is coupled to the top electrodes  124  of transducers  116 ,  130 . Terminal S 4  is coupled to the bottom electrodes  126  of transducers  116 ,  130 . As such, the output of transducer  116  is inverted relative to the input of transducer  114 , and the output of transducer  130  is inverted relative to the input of transducer  128 . The series resonance frequencies of the respective transducers  114 ,  116  as well as respective transducers  128 ,  130  may be the same or different depending on the application. As such, transducers  114  and  116  may have the same or different resonance frequencies. The same applies to transducers  128  and  130 . 
       FIG. 34  illustrates an embodiment where the bottom transducers  86  of different CRFs  132 ,  134  are transversally acoustically coupled. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrode  100  of the top transducer  98  of CRF  132  is coupled to terminal S 1 . The bottom electrode  102  of the top transducer  98  of CRF  132  is coupled to terminal S 2 . The top electrode  100  of the top transducer  98  of CRF  134  is coupled to terminal S 4 . The bottom electrode  102  of the top transducer  98  of CRF  134  is coupled to terminal S 3 . The top electrodes  88  of the bottom transducers  86  are coupled to one another, and the bottom electrodes  90  of the bottom transducers  86  are coupled to one another. 
     In this embodiment, the bottom transducers  86  of CRFs  132 ,  134  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  132  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  134  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     For each CRF  132 ,  134 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may be different between the CRFs  132 ,  134 . For example, the top and bottom transducers  98 ,  86  for CRF  132  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  134  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . In another embodiment, the top and bottom transducers  98 ,  86  for both CRFs  132 ,  134  may have a series resonance frequency of FS 1 . Alternatively, the top transducer  98  for CRF  132  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  132  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  134  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  134  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
       FIG. 35  illustrates an embodiment where transducers  86  of different CRFs  132 ,  134  are transversally acoustically coupled. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrode  100  of the top transducer  98  of CRF  132  is coupled to terminal S 1 . The bottom electrode  102  of the top transducer  98  of CRF  132  is coupled to terminal S 2 . The top electrode  100  of the top transducer  98  of CRF  134  is coupled to terminal S 3 . The bottom electrode  102  of the top transducer  98  of CRF  134  is coupled to terminal S 4 . The top electrode  88  of the bottom transducer  86  of CRF  132  is coupled to the bottom electrode  90  of the bottom transducer  86  of CRF  134 . The bottom electrode  90  of bottom transducer  86  of CRF  132  is coupled to the top electrode  88  of the bottom transducer  86  of CRF  134 . 
     In this embodiment, the bottom transducers  86  of CRFs  132 ,  134  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  132  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  134  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     For each CRF  132 ,  134 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may be different between the CRFs  132 ,  134 . For example, the top and bottom transducers  98 ,  86  for CRF  132  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  134  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . In another embodiment, the top and bottom transducers  98 ,  86  for both CRFs  132 ,  134  may have a series resonance frequency of FS 1 . Alternatively, the top transducer  98  for CRF  132  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  132  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  134  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  134  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
       FIG. 36  illustrates an embodiment where the bottom transducers  86  of two CRFs  132 ,  134  are transversally acoustically coupled and the bottom transducers  86  of two CRFs  138 , 140  are transversally acoustically coupled. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrodes  100  of the top transducers  98  of CRFs  132 ,  138  are coupled to terminal S 1 . The bottom electrodes  102  of the top transducers  98  of CRF  132 ,  138  are coupled to terminal S 2 . The top electrodes  100  of the top transducers  98  of CRF  134 ,  140  are coupled to terminal S 4 . The bottom electrodes  102  of the top transducers  98  of CRF  134 ,  140  are coupled to terminal S 3 . The top electrodes  88  of the bottom transducers  86  of CRFs  132 ,  134  are coupled to one another, and the bottom electrodes  90  of the bottom transducers  86  of CRFs  132 ,  134  are coupled to one another. The top electrodes  88  of the bottom transducers  86  of CRFs  138 ,  140  are coupled to one another, and the bottom electrodes  90  of the bottom transducers  86  of CRFs  138 ,  140  are coupled to one another. 
     In this embodiment, the bottom transducers  86  of CRFs  132 ,  134  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  132  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  134  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     Similarly, the bottom transducers  86  of CRFs  138 ,  140  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  138  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  140  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     As with previous embodiments, the series resonance frequencies for the various transducers  86 ,  98  of the CRFs  132 ,  134 ,  138 ,  140  may be the same or different, as well as combinations of same and different series resonance frequencies. For each CRF  132 ,  134 ,  138 ,  140  depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may be different between the CRFs  132 ,  134 ,  138 ,  140 . For example, the top and bottom transducers  98 ,  86  for CRF  132  may have a series resonance frequency of FS 1 , the top and bottom transducers  98 ,  86  for CRF  134  may have a series resonance frequency of FS 2 , the top and bottom transducers  98 ,  86  for CRF  138  may have a series resonance frequency of FS 3 , and the top and bottom transducers  98 ,  86  for CRF  140  may have a series resonance frequency of FS 4 , where FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. Alternatively, the top transducer  98  for one or more of the CRFs  132 ,  134 ,  138 ,  140  may have a different series resonance frequency. 
       FIG. 37  illustrates an embodiment where transducers  86  of different CRFs  132 ,  134  are transversally acoustically coupled. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrode  100  of the top transducer  98  of CRF  132  is coupled to terminal S 1 . The bottom electrode  102  of the top transducer  98  of CRF  132  is coupled to terminal S 2 . The top electrode  100  of the top transducer  98  of CRF  134  is coupled to terminal S 3 . The bottom electrode  102  of the top transducer  98  of CRF  134  is coupled to terminal S 4 . The top electrode  88  of the bottom transducer  86  of CRF  132  is coupled to the bottom electrode  102  of the top transducer  98  of CRF  134 . The bottom electrode  102  of the top transducer  98  of CRF  132  is coupled to the top electrode  88  of the bottom transducer  86  of CRF  134 . 
     In this embodiment, the bottom transducers  86  of CRFs  132 ,  134  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  132  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  134  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     For each CRF  132 ,  134 , depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may be different between the CRFs  132 ,  134 . For example, the resonance frequencies for the top and bottom transducers  98 ,  86  for both CRFs  132 ,  134  may all be the same. In another embodiment, the top and bottom transducers  98 ,  86  for CRF  132  may have a series resonance frequency of FS 1 , and the top and bottom transducers  98 ,  86  for CRF  134  may have a series resonance frequency of FS 2 , where FS 1  is different than FS 2 . In another embodiment, the top and bottom transducers  98 ,  86  for both CRFs  132 ,  134  may have a series resonance frequency of FS 1 . Alternatively, the top transducer  98  for CRF  132  may have a series resonance frequency of FS 1 , the bottom transducer  86  for CRF  132  may have a series resonance frequency of FS 2 , the top transducer  98  for CRF  134  may have a series resonance frequency of FS 3 , and the bottom transducer  86  for CRF  134  may have a series resonance frequency of FS 4 , wherein FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. 
       FIG. 38  illustrates an embodiment where the bottom transducers  86  of two CRFs  132 ,  134  are transversally acoustically coupled and the bottom transducers  86  of two CRFs  138 , 140  are transversally acoustically coupled. An input signal VP is provided at terminals S 1  and S 2 , and an output signal O/P is provided at terminals S 3  and S 4 . One or more BAW resonators B 5 , B 6  are provided in series between terminals S 1  and S 3  and parallel with one another. As illustrated, terminal S 2  is coupled to a first signal ground, and terminal S 4  is coupled to a second signal ground. 
     In particular, the top electrodes  100  of the top transducers  98  of CRFs  132 ,  138  are coupled to terminal S 1 . The bottom electrodes  102  of the top transducers  98  of CRF  132 ,  138  are coupled to terminal S 2 . The top electrodes  100  of the top transducers  98  of CRF  134 ,  140  are coupled to terminal S 3 . The bottom electrodes  102  of the top transducers  98  of CRF  134 ,  140  are coupled to terminal S 4 . The top electrodes  88  of the bottom transducers  86  of CRFs  132 ,  134  are coupled to one another, and the bottom electrodes  90  of the bottom transducers  86  of CRFs  132 ,  134  are coupled to one another. The top electrodes  88  of the bottom transducers  86  of CRFs  138 ,  140  are coupled to one another, and the bottom electrodes  90  of the bottom transducers  86  of CRFs  138 ,  140  are coupled to one another. 
     In this embodiment, the bottom transducers  86  of CRFs  132 ,  134  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  132  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  134  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     Similarly, the bottom transducers  86  of CRFs  138 ,  140  are transversally acoustically coupled via a common piezoelectric layer  136 . The top transducers  98  are not transversally acoustically coupled. In particular, the bottom transducer  86  of CRF  138  includes a portion (leftmost portion) of a piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . The bottom transducer  86  of CRF  140  includes a portion (rightmost portion) of the same piezoelectric layer  136  sandwiched between a top electrode  88  and a bottom electrode  90 . 
     As with previous embodiments, the series resonance frequencies for the various transducers  86 ,  98  of the CRFs  132 ,  134 ,  138 ,  140  may be the same or different, as well as combinations of same and different series resonance frequencies. For each CRF  132 ,  134 ,  138 ,  140  depending on the characteristics desired of the filter circuit, the series resonance frequencies of the top and bottom transducers  98 ,  86  may be the same or different. The series resonance frequencies for the top and bottom transducers  98 ,  86  may be different between the CRFs  132 ,  134 ,  138 ,  140 . For example, the top and bottom transducers  98 ,  86  for CRF  132  may have a series resonance frequency of FS 1 , the top and bottom transducers  98 ,  86  for CRF  134  may have a series resonance frequency of FS 2 , the top and bottom transducers  98 ,  86  for CRF  138  may have a series resonance frequency of FS 3 , and the top and bottom transducers  98 ,  86  for CRF  140  may have a series resonance frequency of FS 4 , where FS 1 , FS 2 , FS 3 , and FS 4  are unique series resonance frequencies. Alternatively, the top transducer  98  for one or more of the CRFs  80 ,  106 ,  110 ,  112  may have a different series resonance frequency. 
     A variant on the embodiment of  FIG. 38  is provided in  FIG. 39 . The difference is that the top electrode  88  of the bottom transducer  86  of CRF  138  is coupled to the bottom electrode  90  of the bottom transducer  86  of CRF  140 . Further, the bottom electrode  90  of the bottom transducer  86  of CRF  138  is coupled to the top electrode  88  of the bottom transducer  86  of CRF  140 . As such, the location of inversion may vary from one embodiment to another, and certain embodiments need not provide an inversion depending on the application. For any of the embodiments described above, the series resonance frequencies of the BAW resonators B 5 , B 6  may be different from that for the CRFs  80 , etc. 
     While the concepts disclosed herein are described in association with a mobile terminal, these concepts are applicable to any type of communication device that employs wireless communications. Those skilled in the art will recognize numerous modifications and other embodiments that incorporate the concepts described herein. These modifications and embodiments are considered to be within scope of the teachings provided herein and the claims that follow.