Patent Publication Number: US-9405305-B1

Title: Precision voltage reference circuit with tunable resistance

Description:
BACKGROUND 
     The field of the disclosure relates generally to precision voltage reference circuits, and more specifically to a precision voltage reference circuit having a tunable switched capacitor resistance. 
     Many electrical systems, including, for example, control systems and measurement devices, rely on a voltage reference for some aspect of operation. In certain applications, such as long-range guided vehicles, a precision voltage reference (PVR) is critical, because even small shifts in the voltage reference translate to errors in acceleration, position, and rotation. Some vehicles, such as intercontinental missiles and space vehicles, for example, use inertial pendulum-based navigation systems, gyroscopic based navigation systems, or some combination of both to satisfy their low tolerance for error in precision and accuracy. Such systems often require a PVR with stability on the order of 1 part-per-million (ppm) over age, temperature variation, and radiation events. 
     BRIEF DESCRIPTION 
     According to one aspect of the present disclosure, a voltage reference circuit is provided. A voltage reference circuit includes a bridge circuit having a first branch, a second branch, and an amplifier. The bridge circuit is coupled between a precision voltage reference (PVR) node and a ground node. The first branch includes a first resistor of value R 1  coupled to a reference resistor of value Rref at a first intermediate node. The second branch includes a second resistor of value R 1  coupled to a variable resistor of value Rvar at a second intermediate node. Rvar is non-linearly tunable based on the PVR. The amplifier includes a positive input terminal coupled to the second intermediate node and a negative input terminal coupled to the first intermediate node. The amplifier is configured to generate the PVR. 
     According to another aspect of the present disclosure, a method of generating a precision voltage reference (PVR) is provided. The method includes generating a startup voltage for a bridge circuit coupled between the PVR and ground. The method further includes comparing voltages at intermediate nodes of a first branch and a second branch of the bridge circuit to generate the PVR. The method further includes tuning a switched capacitor resistor in the second branch using at least one of a variable frequency control signal and a variable capacitance. The tuning is based on the PVR. 
     The features, functions, and advantages that have been discussed can be achieved independently in various embodiments or may be combined in yet other embodiments further details of which can be seen with reference to the following description and drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic diagram of one embodiment of a voltage reference circuit; 
         FIG. 2  is a schematic diagram of another embodiment of a voltage reference circuit; 
         FIG. 3  is a schematic diagram of yet another embodiment of a voltage reference circuit; 
         FIG. 4  is a schematic diagram of yet another embodiment of a voltage reference circuit; 
         FIG. 5  is a schematic diagram of one embodiment of a switched capacitor resistor; 
         FIG. 6  is a schematic diagram of one embodiment of a varactor; 
         FIG. 7  is a schematic diagram of another embodiment of a varactor; 
         FIG. 8  is a plot of resistance and frequency as a function of a reference voltage for one embodiment of voltage reference circuit; 
         FIG. 9  is a plot of resistance and capacitance as a function of a reference voltage for one embodiment of a voltage reference circuit; 
         FIG. 10  is a plot of intermediate node voltages for one embodiment of a voltage reference circuit; and 
         FIG. 11  is a flow diagram of one embodiment of a method of generating a precision voltage reference. 
     
    
    
     DETAILED DESCRIPTION 
     As used herein, an element or step recited in the singular and preceded by the word “a” or “an” should be understood as not excluding plural elements or steps unless such exclusion is explicitly recited. Furthermore, references to “one embodiment” of the present invention or the “exemplary embodiment” are not intended to be interpreted as excluding the existence of additional embodiments that also incorporate the recited features. 
     In some systems, a PVR source is realized through mechanical apparatuses that are typically bulky, low-precision, and consume a lot of power. Other systems use electronic bandgap reference circuits that are typically noisy and sensitive to radiation. Bandgap reference circuits generally include forward-biased semiconductor p-n junctions that are known to shift with exposure to radiation. Such bandgap reference circuits can achieve as low as 5 ppm per degree C. stability in a tightly controlled environment, when complemented with high-order curvature compensation ancillary circuits. However, practically, certain physical phenomena during operation of the bandgap circuit are subject to shifts on the order of 1000 ppm per degree C., which complicates achieving that 5 ppm per degree C. stability upon exogenous events, such as, for example, radiation doses. Certain other applications for PVR, such as signal sensing and pre-conditioning, analog to digital conversion, digital to analog conversion, battery supervision, and laser diode drivers, for example, can achieve 1 ppm per degree C. using low-voltage circuits and special tunnel diodes, but use costly non-standard semiconductor technology and are still sensitive to radiation. It is realized herein that the precision and reliability of existing PVR circuits is desirable. It is further realized herein that a PVR circuit can be constructed that is temperature and radiation stable to at least 1 ppm using a tunable resistance. 
     Thus, exemplary embodiments may provide a voltage reference circuit configured to generate a PVR that is temperature- and radiation-stable. More specifically, exemplary voltage reference circuits include a bridge circuit that includes an amplifier that self-references, which eliminates dependencies of the PVR from the supply. Some exemplary bridge circuits include a variable resistor configured to be monotonically tuned by the PVR. Some exemplary bridge circuits include a switched capacitor resistor that is configured to be tuned by at least one of a variable frequency control signal and a variable capacitance. Embodiments utilizing a variable frequency control signal generate the variable frequency signal using a voltage controlled oscillator (VCO) that is configured to be tuned based on the PVR. Certain VCOs, such as a relaxation VCO and a differential L-C tank VCO, can be implemented to be particularly temperature-stable and radiation-stable. Embodiments utilizing a variable capacitance, or varactor, tune the capacitance based on the PVR. Certain varactors, such as a MOS varactor, a MOSFET capacitor, and a MEMS varactor, can also be implemented to be temperature- and radiation-stable. 
     In certain embodiments, the variable frequency control signal is paired with a fixed capacitance. Such embodiments fall into a first class of embodiments. In other embodiments, a variable capacitance is paired with a fixed frequency control signal generated by a precision clock device, such as a crystal oscillator, which is also temperature and radiation stable. These embodiments fall into a second class of embodiments. In embodiments utilizing both a variable frequency control signal and a variable capacitance, both the variable frequency control signal and the variable capacitance are tunable based on the PVR, which facilitates further compensation of residual instabilities. Such embodiments fall into a third class of embodiments. 
       FIG. 1  is a schematic diagram of one embodiment of a voltage reference circuit  100 . Voltage reference circuit  100  includes a bridge circuit  110  coupled between a reference voltage node, Vref, and ground node. Bridge circuit  110  includes a first branch having a resistor of value R 1  and another resistor of value Rref. In the middle of the first branch, between R 1  and Rref, is a first intermediate node  120 . Bridge circuit  110  also includes a second branch having a resistor of value R 1  and a variable resistor of value Rvar. In the middle of the second branch, between R 1  and Rvar, is a second intermediate node  130 . In certain embodiments, resistors R 1  in the first and second branches, and Rref are precision resistors, making them temperature- and radiation-stable. 
     Bridge circuit  110  also includes an amplifier  140  coupled as a bridge between first intermediate node  120  and second intermediate node  130 . Amplifier  140  includes a negative input terminal coupled to first intermediate node  120  and a positive input terminal coupled to second intermediate node  130 . Amplifier  140  also includes an output terminal coupled to Vref. In certain embodiments, amplifier  140  includes a plurality of metal-oxide semiconductor field effect transistors (MOSFETs), making amplifier  140  temperature-stable and radiation-stable. 
     During operation, Vref is divided by the first branch and the second branch based on values of R 1  and Rref, and R 1  and Rvar, respectively. A voltage Va presents at first intermediate node  120  and a voltage Vb presents at second intermediate node  130 . Amplifier  140  operates as a linear error amplifier and generates Vref, which is fed back to the branches, serving as a self-reference for bridge circuit  110 . Self-referencing of bridge circuit  110  using amplifier  140  eliminates supply dependence and provides a closed-loop convergence once a startup voltage is applied via a startup circuit (not shown). The startup circuit activates the loop, for example, by raising the voltage Vb at second intermediate node  130  upon power-on. Amplifier  140  is supplied by a substantially non-regulated voltage supply and can be implemented with a power supply rejection (PSR) of at least 100 dB. Furthermore, amplifier  140  operates in the forward path of the closed-loop. 
     Voltage reference circuit  100  is tuned based on the values of R 1 , Rref, and Rvar. More specifically, bridge circuit  110  converges on a voltage output, Vref, which is the PVR, based on the value of Rvar relative to Rref. In certain embodiments, the variable resistor of value Rvar is implemented as a switched capacitor resistor with a fixed capacitance C that is alternately charged and discharged through switches controlled by a sinusoidal or square-wave signal having a frequency F. Such embodiments fall into a first class of embodiments. The frequency of the control signal is tuned based on Vref. As frequency increases, Rvar decreases, because the capacitance in the switched capacitor resistor is constant. 
     In other embodiments, the variable resistor of value Rvar is implemented as a varactor controlled by a stable frequency signal. Such embodiments fall into a second class of embodiments. As the capacitance value C of the varactor increases with Vref, Rvar decreases, because the frequency F operating Rvar is constant. The tunability of switched capacitor resistor Rvar, whether through F or C, facilitates tuning of voltage reference circuit  100  to a steady-state condition where Vref is temperature- and radiation-stable. The steady-state Vref is the desired PVR. In alternative embodiments, R 1  is also tunable and may include a switched capacitor resistor similar to Rvar, facilitating further compensation and PVR stabilization. 
     In certain embodiments, frequency F is variable and tunable based on Vref and capacitance C is fixed, which is referred to as the first class of embodiments. In other embodiments, frequency F is stable and capacitance C is tunable based on Vref, which is referred to as the second class of embodiments. In some embodiments, referred to as a third class of embodiments, both frequency F and capacitance C are variable and tunable based on Vref. 
       FIG. 2  is a schematic diagram of another embodiment of a voltage reference circuit  200 . Voltage reference circuit  200  falls into the first class of embodiments. Voltage reference circuit  200  includes a bridge circuit  210  coupled between Vref and ground, similar to bridge circuit  110  (shown in  FIG. 1 ). Bridge circuit  210  includes resistors R 1 , resistor Rref, and switched capacitor resistor Rvar (all shown in  FIG. 1 ). Bridge circuit  210  also includes a first intermediate node  220 , a second intermediate node  230 , and an amplifier  240  coupled between. Bridge circuit  210  operates the same as bridge circuit  110 . 
     Voltage reference circuit  200  further includes a voltage controlled oscillator (VCO)  250 . Switched capacitor resistor Rvar is controlled by a periodic signal of frequency F. The periodic signal is generated by VCO  250  at frequency F. Frequency F is tuned based on Vref and capacitance C is fixed. In such embodiments, VCO  250  can be, for example, a relaxation VCO or a differential LC-tank VCO. Voltage reference circuit  200  does not use an external precision clock. 
       FIG. 3  is a schematic diagram of yet another embodiment of a voltage reference circuit  300 . Voltage reference circuit  300  falls into the second class of embodiments. Voltage reference circuit  300  includes a bridge circuit  302  coupled between Vref′ and ground, similar to bridge circuit  110  (shown in  FIG. 1 ), a phase-lock loop (PLL) circuit  304 , and a summer  306 . Bridge circuit  302  includes resistors R 1 , resistor Rref, and switched capacitor resistor Rvar (all shown in  FIG. 1 ). Bridge circuit  302  also includes a first intermediate node  308 , a second intermediate node  310 , and an amplifier  312  coupled between. Bridge circuit  302  operates the same as bridge circuit  110 . 
     In the switched capacitor resistor, Rvar is implemented as a varactor (not shown) having a variable capacitance C and tunable based on Vref. The switched capacitor resistor is controlled by a periodic signal having a fixed frequency F. The periodic signal is generated by precision clock  314 . Precision clock  314 , in certain embodiments, includes a crystal oscillator for generating the fixed frequency F periodic signal. In certain embodiments, PLL circuit  304  and summer  306  are omitted and precision clock  314  directly drives the switched capacitor resistor Rvar. 
     PLL circuit  304  further includes a phase and frequency detector (PFD)  316 , a low-pass filter  318 , and a VCO  320 . VCO  320  is tuned by a varactor of the same type as in the switched capacitance resistor of value Rvar. VCO  320  is configured to generate a sinusoidal signal that is fed back to PFD  316  where it is compared to the periodic signal of frequency F generated by precision clock  314 . PLL circuit  304  tunes VCO  320  to emit a sinusoidal signal of frequency F. PLL circuit  304  thereby generates an internal tuning voltage Vvco that compensates for any exogenous variations of the varactors in VCO  320  and the switched capacitor resistor, Rvar, including temperature, radiation, and process corner skew, among others. Tuning voltage Vvco is applied by PLL circuit  304  to VCO  320  to substantially counter the same variations that impact the varactor in Rvar. Therefore, when Vvco is added to Vref′ at summer  306 , the resulting voltage, Vref, is compensated for such exogenous effects. 
     In alternative embodiments, tuning voltage Vvco is summed with a voltage Vb at second intermediate node  310  and applied directly to the switched capacitor resistor to tune the varactor of Rvar. In such an embodiment, Vref′ is tuned to Vref. 
       FIG. 4  is a schematic diagram of yet another embodiment of a voltage reference circuit  400 . Voltage reference circuit  400  falls into the third class of embodiments. Voltage reference circuit  400  includes a bridge circuit  402  coupled between a Vref node and a ground node, similar to bridge circuit  110  (shown in  FIG. 1 ). Voltage reference circuit further includes a VCO  404  configured to generate a voltage output Vvco having a frequency F that is tunable based on Vref. 
     Bridge circuit  402  includes resistors R 1 , resistor Rref, and amplifier  406  (all shown in  FIG. 1 ). Bridge circuit  402  also includes a switched capacitor resistor  408  of value Rvar. Bridge circuit  402  also includes a first intermediate node  410  and a second intermediate node  412 . Bridge circuit  402  operates the same as bridge circuit  110 . 
     Switched capacitor resistor  408  includes a varactor  414  having two control terminals respectively coupled to the Vref node through a resistive network  416  and the ground node. Varactor  414  has a capacitance C that is tunable by Vref via the control terminals. Varactor  414  includes a constant capacitance  418  and a semiconductor capacitor  420 . The capacitance of semiconductor capacitor  420  is tunable by the voltage present across the two control terminals, which depends on voltage Vref. Consequently, the resistance, Rvar, of switched capacitor resistor  408  is tunable based on Vref. 
     Switched capacitor resistor  408  also includes a switch  422  and a switch  424 , each having a control terminal coupled to the output of VCO  404 , Vvco. Switch  422  and switch  424 , as controlled by Vvco at frequency F, control the charge that moves through varactor  414 . As described above, the frequency F of the output of VCO  404  is tunable based on Vref. Consequently, the resistance, Rvar, of switched capacitor resistor  408  is further tunable based on Vref. In certain embodiments, switches  422  and  424  are implemented with MOSFET devices, which can be complementary or not depending on how the driving phases are derived from the oscillation of Vvco. For example, Vvco can be used to generate non-overlapped phases that can be used to drive two N-MOSFETs, rather than an inverter comprised of one N-MOSFET and one P-MOSFET. 
     In certain embodiments, VCO  404  includes a varactor of the same type as in switched capacitor resistor  408 . For example, VCO  404  can be implemented as an L-C tank VCO. By using the same type of varactor, the temperature- and radiation-stability of voltage reference circuit  400  are improved, because variations in Rvar due to temperature or radiation events are further compensated for by the temperature and radiation response of VCO  404 . 
       FIG. 5  is a schematic diagram of one embodiment of a switched capacitor resistor  500  for use in a voltage reference circuit, such as voltage reference circuits  100 ,  200 ,  300 , and  400  (shown in  FIGS. 1-4 ). Switched capacitor resistor  500  includes a capacitor  510 , a first MOSFET  520 , and a second MOSFET  530 . First MOSFET  520  and second MOSFET  530  are coupled in series, source to drain, between a first terminal V 1  and a second terminal V 2 . First MOSFET  520  and second MOSFET  530  can be NMOS or PMOS devices. In alternative embodiments, first MOSFET  520  and second MOSFET  530  can be replaced by any other suitable switching device, including, for example, relays. Electromechanical relays have an advantage, for example, that they are both more temperature- and radiation-stable than their semiconductor counterparts. 
     Capacitor  510  is coupled between ground and a node between first MOSFET  520  and second MOSFET  530 . First MOSFET  520  and second MOSFET  530  are respectively controlled by a first switch signal  51  and a second switch signal S 2 , at the respective gates of first MOSFET  520  and second MOSFET  530 . First MOSFET  520  and second MOSFET  530  are opened and closed alternatingly. In certain embodiments, first switch signal  51  and second switch signal S 2  are implemented as a single periodic signal having a frequency F. For example, in an embodiment having complementary NMOS and PMOS switches, a single periodic signal can control both first MOSFET  520  and second MOSFET  530 . 
     When a voltage, V, is presented at V 1 , capacitor  510  is charged when first MOSFET  520  is closed and second MOSFET  530  is open. When first MOSFET  520  opens and second MOSFET  530  closes, capacitor  510  discharges, moving the charge to V 2 , which may be connected to ground, for example. The movement of the charge from V 1  to V 2  is a current. The amount of current is quantified by the change in charge over a change in time, or I=dq/dt, that can be expressed, for a capacitance C and a control signal frequency f, as I=C·V·f. Kirchhoff&#39;s law, R=V/I, permits the resistance of switched capacitor resistor  400  to be expressed as R=1/(C·f). The tunability of capacitance C, frequency f, or both, permits classification of embodiment voltage reference circuits into the first class, the second class, and the third class described above with respect to  FIGS. 2-4 . 
     In certain embodiments, capacitor  510  is a fixed capacitance parallel plate capacitor and first MOSFET  520  and second MOSFET  530  are controlled by a variable frequency signal as first switch signal  51  and second switch signal S 2 . As the variable frequency signal increases in frequency, the resistance of switched capacitor resistor  500  decreases. In certain embodiments, capacitor  510  is a variable capacitance, such as a varactor, and first MOSFET  520  and second MOSFET  530  are controlled by a fixed frequency signal. As the variable capacitance increases, the resistance of switched capacitor resistor  500  decreases. In certain embodiments, capacitor  510  is a variable capacitance, such as a varactor, and first MOSFET  520  and second MOSFET  530  are controlled by a variable frequency signal. Varying both the capacitance of capacitor  510  and the frequency of first switch signal  51  and second switch signal S 2  facilitates finer tuning and compensation of the resistance of switched capacitor resistor  500 . 
     In certain embodiments, capacitor  510  is implemented as a varactor on silicon, such as a silicon junction or MOS capacitor. In other embodiments, capacitor  510  is implemented with discrete components, such as one or more relays controlling a varactor. In certain embodiments, capacitor  510  is a varactor implemented using micro-electromechanical systems (MEMS) to form an electrically controlled parallel plate capacitor. In a MEMS varactor, two terminals are used for controlling the separation of the parallel plates by pushing or pulling the plates together or apart. Two other terminals are used as the terminals of the capacitor. A MEMS varactor provides good temperature and radiation stability, because the dielectric and plates are both stable. The MEMS varactor also requires an externally provided control voltage. 
       FIG. 6  is a schematic diagram of one embodiment of a semiconductor varactor  600  for use in a switched capacitor resistor, such as switched capacitor resistor  500  (shown in  FIG. 5 ) and in a capacitance tuned VCO, such as VCO  250  and VCO  320  (shown in  FIGS. 2 and 3 ). Semiconductor varactor  600  includes a constant capacitor C 1 , a varactor diode D, and a constant capacitor C 2  coupled in series between a voltage V+ and a voltage V−. A first control terminal Vc 1  is coupled to the cathode of varactor diode D through a resistor R 1 . A second control terminal Vc 2  is coupled to the anode of varactor diode D through a resistor R 2 . Semiconductor varactor  600  uses the voltage-dependent capacitance of the reversed-biased p-n junction of varactor diode D to tune to a desired capacitance. The combined effects of voltages V+, V−, and voltages applied at Vc 1  and Vc 2 , facilitate tuning semiconductor varactor  600  by transforming a 2-terminal device in varactor diode D into a 4-terminal device in semiconductor varactor  600 . In alternative embodiments, semiconductor varactor  600  utilizes a MOS capacitor with a voltage-dependent capacitance. 
       FIG. 7  is a schematic diagram of another embodiment of a varactor  700  for use in a switched capacitor resistor, such as switched capacitor resistor  500  (shown in  FIG. 5 ) and in a capacitance tuned VCO, such as VCO  250  and VCO  320  (shown in  FIGS. 2 and 3 ). Varactor  700  includes an MOSFET  710  having a gate terminal G, a drain terminal D, a source terminal S, and a body  720 . MOSFET  710  is wired as a capacitor by coupling source S and drain D to body  720  and providing a body terminal B. The capacitance of varactor  700  is measured across gate terminal G and body terminal B, and depends on the voltage across those terminals. Body  720  can be implemented with a silicon well of the same or opposite polarity as diffusion/implants of source S and drain D, facilitating operation of MOSFET  710  as a varactor or FET capacitor. MOS varactors provide good radiation stability. 
       FIG. 8  is a plot  800  of resistance R and frequency F as functions of a reference voltage Vref for an exemplary voltage reference circuit, such as voltage reference circuits  100 ,  200 , and  400  (shown in  FIGS. 1, 2, and 4 ). More specifically, resistance R is that of a switched capacitor resistor, such as switched capacitor resistor  500  (shown in  FIG. 5 ), and frequency F is that of a sinusoidal signal controlling the switching of first MOSFET  520  and second MOSFET  530 . The sinusoidal signal, in certain embodiments, is generated by an external precision clock device, such as a VCO or capacitance-tuned VCO. 
     Frequency F is tuned monotonically based on voltage Vref. Plot  800  illustrates that F increases linearly with Vref. In alternative embodiments, F may increase non-linearly with Vref. Given the hyperbolic R=1/(C·f) relationship for the switched capacitor resistor, resistance R decreases non-linearly with an increase in Vref. 
       FIG. 9  is a plot  900  of resistance R and capacitance C as a function of a reference voltage Vref for an exemplary voltage reference circuit, such as voltage reference circuits  100 ,  300 , and  400  (shown in  FIGS. 1, 3, and 4 ). More specifically, resistance R is that of a switched capacitor resistor, such as switched capacitor resistor  500  (shown in  FIG. 5 ), and capacitance C is that of a varactor, such as varactor  600  and  700  (shown in  FIGS. 6 and 7 ) for use in switched capacitor resistor  500 . 
     Capacitance C is tuned monotonically based on voltage Vref. Plot  900  illustrates that C increases linearly with Vref. In alternative embodiments, C may increase non-linearly with Vref. Given the hyperbolic R=1/(C·f) relationship for the switched capacitor resistor, resistance R decreases non-linearly with an increase in Vref. 
       FIG. 10  is a plot  1000  of bridge voltages for voltage reference circuit  100 , or for voltage reference circuits  200 ,  300 , and  400  (shown in  FIGS. 1-4 ). For a bridge circuit coupled between Vref and ground, such as bridge circuit  110 , a voltage Va presents across reference resistance Rref at first intermediate node  120 . Va is a result of a constant voltage division of Vref across the first branch having resistance R 1  and Rref in series. Similarly, a voltage Vb presents across a switched capacitor resistor having a resistance of Rvar at second intermediate node  130 . Vb is a result of a variable voltage division of Vref across the second branch having a resistance R 1  and Rvar in series. 
     Plot  1000  illustrates that voltage Va across Rref increases linearly with Vref. Plot  1000  also illustrates that voltage Vb across Rvar increases non-linearly with Vref. Plots  800  and  900  illustrate the resistance of a switched capacitor resistor varies inversely and non-linearly with capacitance and frequency. In voltage reference circuit  100 , frequency, capacitance, or both are tuned based on Vref. Switched capacitor resistor Rvar in bridge circuit  110  likewise varies inversely and non-linearly with Vref. Voltage Vb can therefore be expressed as: 
             Vb   =       Vref   ·     R   ⁡     (   Vref   )             R   ⁢           ⁢   1     +     R   ⁡     (   Vref   )                 
The variation of Vb with decreasing values of R(Vref) diminishes in the segment of Vb illustrated in plot  1000 . Amplifier  140  causes bridge circuit  110  to balance voltages Va and Vb, and voltage reference circuit  100  to converge on a single, non-trivial stable Vref, referred to as a PVR, which is illustrated as the intersection of Va and Vb. Convergence on the trivial zero solution is avoided by using a startup circuit to drive the loop of bridge circuit  110  to converge on the non-trivial stable PVR.
 
       FIG. 11  is a flow diagram of one embodiment of a method  1100  of generating a precision voltage reference. Method  1100  begins at a start step  1110 . At a startup step  1120 , a startup voltage is applied to a bridge circuit coupled between a Vref node and a ground node, such as bridge circuit  110  (shown in  FIG. 1 ). At a comparing step  1130 , voltages at intermediate nodes of the two branches of the bridge circuit are compared, generating a resulting PVR at the Vref node. 
     At a tuning step  1140 , the PVR is used to tune a switched capacitor resistor in the second branch of the bridge circuit. The switched capacitor resistor is tunable by at least one of a variable frequency control signal and a variable capacitance. The tuned resistance of the switched capacitor resistor operates to tune the bridge circuit to the desired PVR. The method ends at an end step  1150 . 
     This written description uses examples to disclose various embodiments, which include the best mode, to enable any person skilled in the art to practice those embodiments, including making and using any devices or systems and performing any incorporated methods. The patentable scope is defined by the claims, and may include other examples that occur to those skilled in the art. Such other examples are intended to be within the scope of the claims if they have structural elements that do not differ from the literal language of the claims, or if they include equivalent structural elements with insubstantial differences from the literal languages of the claims.