Patent Publication Number: US-7898187-B1

Title: Circuit and method for average-current regulation of light emitting diodes

Description:
TECHNICAL FIELD 
     This disclosure is generally directed to regulators for light emitting diodes and more specifically to a circuit and method for average-current regulation of light emitting diodes. 
     BACKGROUND 
     Many different devices use light emitting diodes (LEDs), such as flashlights, traffic control signals, flat panel displays, mobile telephone displays, vehicle taillights, and light bulbs. The LEDs are typically current driven devices, meaning the LEDs are controlled by controlling the amount of current provided to the LEDs. Ideally, the current supplied to one or more LEDs is controlled at a minimal cost. 
     LED control has traditionally been done using floating buck regulators in peak-current mode, meaning the regulators regulate the peak current provided to the LEDs at switch-on times. In contrast, the average current used by the LEDs may be very different from the LEDs&#39; peak current, and loosely controlled parameters (such as inductance and switching frequency) may alter the relation between the LEDs&#39; average current and peak current. As a result, it is often not possible to predict or control the LEDs&#39; average current based on peak current levels. 
     Another technique that has been devised for controlling LEDs involves modulating a reference voltage used to regulate the LEDs. However, this technique often requires using a matching resistor-capacitor (RC) filter to remove alternating current (AC) modulation. The RC filter may introduce delay into the control loop, and it may limit the range of switching frequencies that can be achieved. For example, at too low of a switching frequency, the RC filter may be unable to remove the AC components. At a higher frequency, the delay introduced by the RC filter may impede the bandwidth that can otherwise be achieved. 
       FIGS. 1 and 2  illustrate conventional regulators for regulating LEDs. As shown in  FIG. 1 , a circuit  100  includes a conventional floating buck converter  102 , a voltage comparator  104 , a leading edge blanking unit  106 , and pulse width modulation (PWM) control logic  108 . The converter  102  includes one or more LEDs  110  coupled in series with one another. The LEDs  110  are also coupled in parallel with a capacitor  112  and a diode  114 . An inductor  116  is coupled between the LEDs  110  and the diode  114 . The diode  114  and the inductor  116  are coupled to a transistor  118 , which is also coupled to a resistor  120 . The gate of the transistor  118  is coupled to a gate driver  122 . 
     The signal provided by the gate driver  122  to the transistor  118  is based on the operation of the voltage comparator  104 , the leading edge blanking unit  106 , and the PWM control logic  108 . The voltage comparator  104  compares an output of the leading edge blanking unit  106  to a reference voltage V REF . An output of the voltage comparator  104  is provided to the PWM control logic  108 , which includes two pulse generators  124 - 126  and three NOR gates  128 - 132 . Each of the pulse generators  124 - 126  receives an input signal having approximately a 50% duty cycle. The pulse generators  124 - 126  generate signals having pulses with approximately the same switching period as the input signal. However, the pulses generated by the pulse generator  124  are shorter than the pulses generated by the pulse generator  126 . The outputs of the voltage regulator  104  and the pulse generators  124 - 126  are received by the NOR gates  128 - 132 . The NOR gate  132  produces an output signal that is provided to the gate driver  122  for use in controlling the transistor  118 , where the output signal produced by the NOR gate  132  has a desired duty cycle. 
     The leading edge blanking unit  106  receives the voltage produced between the transistor  118  and the resistor  120 . The leading edge blanking unit  106  removes a spike in that voltage at the beginning of each switching period to provide a smoother input to the voltage comparator  104 . The spike could represent switching noise generated by switching the transistor  118 . 
     In this conventional circuit  100 , no information regarding the current through the inductor  116  is available when the transistor  118  is switched off. As a result, a sensed voltage SEN (the voltage across the resistor  120 ) is not related to the average current through the inductor  116 . A peak-current regulator (formed from components  104 - 108 ) turns the transistor  118  off when the sensed voltage SEN is above the reference voltage V REF . However, this is very different from regulating the average current through the LEDs  110 . 
     Another convention approach is shown in  FIG. 2 , where a circuit  200  modulates a reference voltage V REF  used to regulate the LEDs  110 . The circuit  200  includes the floating buck converter  102 , voltage comparator  104 , and PWM control logic  108  described above. The circuit  200  also includes an operational transconductance amplifier  202 , transistors  204 - 210 , a voltage source  212 , resistors  214 - 218 , capacitors  220 - 224 , and an inverter  226 . The gate of the transistor  204  is coupled to an input of the inverter  226 , the gate of the transistor  208 , and the output of the gate driver  122 . The transistor  204  is also coupled between the transistor  118  and the resistor  120 . The gate of the transistor  206  is coupled to the output of the inverter  226 . 
     In this circuit  200 , the signal generated between the transistors  208 - 210  is generally a square wave with a maximum voltage of V REF . The signal generated between the transistors  204 - 206  has peaks that are greater than the voltage V REF . The transistors  204 - 206 , resistor  214 , and capacitor  220  produce a signal supplied to one input of the operational transconductance amplifier  202 . The transistors  208 - 210 , voltage source  212 , resistor  216 , capacitor  222 , and inverter  226  produce a signal supplied to another input of the operational transconductance amplifier  202 . Both inputs to the operational transconductance amplifier  202  are generally triangular waves. The output of the operational transconductance amplifier  202  is supplied to the resistor  218  and capacitor  224 , as well as to the voltage comparator  104 . The other input to the voltage comparator  104  is a sawtooth voltage signal having approximately the same switching period as the signals in the PWM control logic  108 . 
     In this approach, the reference voltage V REF  is switched on and off and may have the exact same duty factor as the transistor  118 . Several RC filters with a time constant much greater than the switching period are used to remove AC components in order to extract the average current and voltage. This, however, leads to large space requirements and limited switching frequency ranges. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of this disclosure and its features, reference is now made to the following description, taken in conjunction with the accompanying drawings, in which: 
         FIGS. 1 and 2  illustrate conventional regulators for regulating LEDs; 
         FIG. 3  illustrates an example regulator for regulating LEDs according to this disclosure; and 
         FIGS. 4 through 6  illustrate example operating characteristics of the regulator shown in  FIG. 3 . 
     
    
    
     DETAILED DESCRIPTION 
       FIGS. 3 through 6 , discussed below, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the invention may be implemented in any type of suitably arranged device or system. 
       FIG. 3  illustrates an example regulator  300  for regulating LEDs according to this disclosure. The embodiment of the regulator  300  shown in  FIG. 3  is for illustration only. Other embodiments of the regulator  300  could be used without departing from the scope of this disclosure. 
     As shown in  FIG. 3 , the regulator  300  is used to drive one or more LEDs  302 , which are coupled in parallel with a diode  304  and a capacitor  306 . An inductor  308  is coupled between the LEDs  302  and the diode  304 . A voltage source  310  provides the input voltage V IN  for the LEDs  302 . In particular embodiments, the capacitor  306  represents a 10 μF capacitor, the inductor  308  represents a 10 μH inductor, and the voltage source  310  represents a 12 VDC source. 
     In this example embodiment, the regulator  300  regulates the illumination of the LEDs  302  by controlling whether there is a path from the LEDs  302  to ground via a resistor  312 . In other words, the regulator  300  may act as a switch. When the regulator  300  conducts, a path exists from the LEDs  302  to ground through the resistor  312 . When the regulator  300  does not conduct, no path exists from the LEDs  302  to ground through the resistor  312 . In this way, the regulator  300  can control the illumination produced by the LEDs  302 . In particular embodiments, the resistor  312  represents a 1Ω resistor. 
     In this example, the regulator  300  includes a voltage source  314 , which produces a reference voltage V REF . The reference voltage V REF  may ramp up during an initial period before reaching a desired value. In particular embodiments, the voltage source  314  represents a 1V source. 
     The reference voltage V REF  is provided as one input to an error amplifier  316 . The error amplifier  316  compares the reference voltage V REF  against a sensed voltage (SEN). The sensed voltage SEN represents the voltage across the resistor  312 . The error amplifier  316  generates a current based on whether the reference voltage V REF  exceeds the sensed voltage SEN. The output of the error amplifier  316  is coupled to a voltage source  318  and a capacitor  320 . In particular embodiments, the voltage source  318  represents a 500 mV source, and the capacitor  320  represents a 10 pF capacitor. 
     The voltage source  318  and a voltage source  322  are coupled as inputs to a PWM comparator  324 . The voltage source  322  generates a sawtooth voltage V RAMP . The voltage V RAMP  is inverted by the PWM comparator  324  or an external inverter. In particular embodiments, the voltage source  322  represents a 2V source. 
     The PWM comparator  324  compares its inputs and generates an output PWM based on the comparison. The PWM comparator  324  provides the desired pulse width modulation in the driving of the LEDs  302 . The output of the PWM comparator  324  is provided to a gate driver  326 , which drives the gate of a transistor  328 . The transistor  328  acts as a switch to either allow or block a path from the LEDs  302  to ground via the resistor  312 . 
     In this example, the regulator  300  also includes two inverters  330 - 332 , an AND gate  334 , a voltage source  336 , and a NOR gate  338 . The inverters  330 - 332  operate to detect if the transistor  328  is turned on or off by sensing the transistor&#39;s gate voltage. The AND gate  334  performs a logical AND operation of the output of the inverter  332  and the PWM signal. The output of the AND gate  334  is provided to the NOR gate  338 , along with a clock signal CLK 2  from the voltage source  336 . In particular embodiments, the voltage source  336  represents a 5V source. The output of the NOR gate  338  enables or disables the error amplifier  316 . 
     In one aspect of operation, the reference voltage V REF  can be kept approximately constant, and the error amplifier  316  (when turned on) may continuously compare the reference voltage V REF  and the sensed voltage SEN. However, the error amplifier  316  can be turned on only when there is an active sensed voltage SEN. When the transistor  328  is off or non-conductive, the error amplifier  316  can be turned off as well. The output of the error amplifier  316 , which controls the duty factor of the PWM signal, may be held constant when the error amplifier  316  is off. Turning the error amplifier  316  off may lower power consumption of the regulator  300 . When the transistor  328  is on or conductive, the error amplifier  316  is on and integrates the error voltage (SEN−V REF ) over the period when the transistor  328  is on. 
     The error amplifier  316  can be turned on and off by controlling its supply using the inverters  330 - 332 , AND gate  334 , voltage source  336 , and NOR gate  338 . For example, the output of the NOR gate  338  can turn the error amplifier  316  on just after the transistor  328  becomes conductive. The output of the NOR gate  338  can also turn the error amplifier  316  off before the transistor  328  becomes non-conductive. This may help to reduce or eliminate the possibility of noise affecting the output of the error amplifier  316 . This noise could, for example, come from switching the transistor  328  or from changing the state of the gate driver  326 . In this example, the clock signal CLK 2  is used to routinely turn the error amplifier  316  on even when the duty cycle of PWM is zero. This can be done to prevent the error amplifier  316  from being turned completely off for an extended period of time. 
     In a steady state, any change in the charge of the capacitor  320  may be zero over each cycle. If the current in the inductor  308  is linear, the point where SEN=V REF  may represent the midpoint of the up-ramp (and the down-ramp) of the inductor current. This is also the average inductor current. As a result, the duration that the output of the error amplifier  316  is positive may equal the duration that the output of the error amplifier  316  is negative. 
     As noted above, noise can be reduced in the output of the error amplifier  316 . The regulator  300  can also reduce noise in other ways. For example, noise may have no net direct current (DC) term and may integrate to zero over time. Because of this, it may not change the average current through the inductor  308 . Conventional techniques require elaborate blanking schemes to minimize the effects of switching noise when switching a transistor on and off (see the description of  FIG. 1  above). The regulator  300  need not use elaborate blanking schemes to minimize the effects of switching noise. 
     As shown in  FIG. 3 , the error amplifier  316  and the PWM comparator  324  operate using bias currents BIAS 1  and BIAS 2 .  FIG. 3  illustrates one example mechanism for generating the bias currents. In this example, a voltage source  340  is coupled to the bases of two transistors  342 - 344 . The transistor  342  is coupled in series with a resistor  346 , and the transistor  344  is coupled in series with a resistor  348 . The bias currents are generated at the collectors of the transistors  342 - 344 . In particular embodiments, the voltage source  340  represents a 1.25 VDC source, the transistors  342 - 344  represent NPN bipolar transistors, and the resistors  346 - 348  represent 54.8 kΩ resistors. In addition, various components in the regulator  300  (such as the error amplifier  316  and the logic gates) may derive their operating power from a supply voltage VDD provided by a voltage source  350 . In particular embodiments, the voltage source  350  represents a 5V source. The relationships between the various signals generated in the regulator  300  are also shown in  FIG. 3 . 
     Each of the components of the regulator  300  could be implemented in any suitable hardware, software, firmware, or combination thereof. For example, the error amplifier  316  could be implemented in hardware as a difference amplifier or as a comparator and a charge pump. Also, the regulator  300  as a whole could be implemented using digital circuitry, software, or any other suitable technology. 
     The regulator  300  may provide various advantages depending on its implementation. For example, the regulator  300  may require no RC filters to operate, and the associated limitation in switching frequency may not be present in the regulator  300 . Also, the regulator  300  may be smaller than conventional LED regulators, reducing or minimizing the size and cost of the regulator  300 . The smaller area occupied by the regulator  300  may allow the regulator  300  to be packaged in a smaller form factor. In addition, the regulator  300  may require no additional input or output pins compared to convention regulators. Other or additional benefits could also be obtained with the regulator  300 , depending on the specific implementation used. 
     Although  FIG. 3  illustrates one example of a regulator  300  for regulating LEDs, various changes may be made to  FIG. 3 . For example, the error amplifier  316  could be kept on at all times, and the output of the error amplifier  316  could be switched into and out of the circuit using a transistor switch. As another example, any other arrangement of logic gates or other structures that provide part or all of the functionality of  FIG. 3  could be used. 
       FIGS. 4 through 6  illustrate example operating characteristics of the regulator  300  shown in  FIG. 3 . The operating characteristics of the regulator  300  shown in  FIGS. 4  through  6  are for illustration and explanation only. The regulator  300  could operate in any other or additional manner without departing from the scope of this disclosure. 
       FIG. 4  represents simulation results obtained using the regulator  300  when the LEDs  302  are turned on and the inductor current ramps up towards its specific voltage. In  FIG. 4 , line  402  represents the reference voltage V REF  produced by the voltage source  314 . In this example, the reference voltage V REF  increases from 0V to 1V within 1 ms. Line  404  represents the current through the inductor  308 , which shows a corresponding increase from 0 A to 1 A in approximately the same time period. In other words, the average inductor current tracks the reference voltage V REF . 
       FIG. 5  illustrates the operation of the regulator  300  during one cycle. Line  502  in  FIG. 5  represents the operation of the error amplifier  316 , where the error amplifier  316  is turned on when the signal represented by line  502  is at 5V and turned off when the signal is at 0V. Line  504  represents the reference voltage V REF , and line  506  represents the sensed voltage SEN and the inductor current. The reference voltage V REF  is generally constant at 1V. The sensed voltage SEN and the inductor current approximately match and increase/decrease together. In this example, the sensed voltage SEN follows the inductor current during ramp up (when the transistor  328  is on) and falls to ground during ramp down (when the transistor  328  is off). However, the average inductor current stays close to V REF . As shown here, the regulator  300  is able to control the average current through the LEDs  302 . 
       FIG. 6  illustrates the operation of the regulator  300  during one cycle in a similar way as is done in  FIG. 5 . In  FIG. 6 , a parasitic inductance of 20 nH has been placed in series with the resistor  312 . Line  602  represents the reference voltage V REF , and line  604  represents the inductor current and the sensed voltage SEN. This shows the regulator  300  may operate properly even in the presence of parasitic elements. 
     Although  FIGS. 4 through 6  illustrate examples of the operating characteristics of the regulator  300  shown in  FIG. 3 , various changes may be made to  FIGS. 4 through 6 . For example, the regulator  300  could operate in any other or additional manner. The graphs shown in  FIGS. 4 through 6  are simply provided for illustration and explanation only and do not limit the operations that could be performed by the regulator  300 . 
     In some embodiments, various functions described above can be implemented or supported by a computer program that is formed from computer readable program code and that is embodied in a computer readable medium. The phrase “computer readable program code” includes any type of computer code, including source code, object code, and executable code. The phrase “computer readable medium” includes any type of medium capable of being accessed by a computer, such as read only memory (ROM), random access memory (RAM), a hard disk drive, a compact disc (CD), a digital video disc (DVD), or any other type of memory. 
     It may be advantageous to set forth definitions of certain words and phrases that have been used within this patent document. The term “couple” and its derivatives refer to any direct or indirect communication between two or more components, whether or not those components are in physical contact with one another. The terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation. The term “or” is inclusive, meaning and/or. The phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like. 
     While this disclosure has described certain embodiments and generally associated methods, alterations and permutations of these embodiments and methods will be apparent to those skilled in the art. Accordingly, the above description of example embodiments does not define or constrain this invention. Other changes, substitutions, and alterations are also possible without departing from the spirit and scope of this invention as defined by the following claims.