Patent Publication Number: US-8525584-B2

Title: Automatic cutoff frequency adjusting circuit and portable digital assistant

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present application claims priority from Japanese patent application JP 2010-157703 filed on Jul. 12, 2010, the content of which is hereby incorporated by reference into this application. 
     BACKGROUND 
     The present invention relates to an automatic filter cutoff frequency adjusting technique, and particularly to a technique effective when applied to an automatic filter cutoff frequency adjusting circuit incorporated in, for example, a portable digital assistant. 
     Conventionally, a radio signal processing circuit is configured with discrete components of the respective functional blocks (an amplifier for amplifying a signal, a mixer for converting the frequency of the signal, a filter for passing only a desired band of the signal, and the like). However, recent semiconductor technology improvements have made it possible to incorporate the functional blocks configuring the radio signal processing circuit into one or more semiconductor chips. The radio signal processing circuit incorporated in the semiconductor chip converts a high-frequency signal received from an antenna into a signal of a lower frequency band with high quality (low noise, suppression of the signals of bands other than a desired signal band, and the like). 
     To achieve the radio signal processing circuit at low cost, it is necessary to incorporate more functional blocks configuring the radio signal processing circuit into one semiconductor chip. One of the obstacles to this objective is to incorporate a filter circuit for suppressing the signals of the undesired bands into the semiconductor chip. A SAW (Surface Acoustic Wave) filter, a dielectric filter, or the like is used as the filter circuit to suppress signals present in the bands other than the desired band. However, it is not possible to incorporate the SAW filter or the dielectric filter into the semiconductor chip. 
     In general, the radio signal processing circuit of discrete components is configured with a superheterodyne system which requires the SAW filter or the dielectric filter (see e.g. Non-patent Document 1 for the superheterodyne system); however, it is not possible to incorporate the SAW filter or the dielectric filter into the semiconductor chip. Accordingly, in the case where the radio signal processing circuit made of semiconductors is configured with the superheterodyne system, the SAW filter or the dielectric filter is provided outside the semiconductor chip, which increases the number of components and the mounting area. 
     There is proposed a radio signal processing circuit system that does not require the SAW filter or the dielectric filter, utilizing a feature of the semiconductor circuit that absolute component values vary among semiconductor chips whereas relative component values within one semiconductor chip coincide with each other with high accuracy. This system is a zero-IF system, a low-IF system, or the like, which does not require an external SAW filter or dielectric filter and suppresses signals present in the bands other than the desired band through the use of filters that can be incorporated into the semiconductor chip (it may be necessary that some filters be provided outside according to a radio system or system requirements). 
     In the zero-IF system and the low-IF system, a channel filter for eliminating signals other than a desired channel is disposed in a stage for processing the signal of a low frequency band after frequency conversion by a mixer circuit. By disposing the filter in the stage for processing the signal of the low frequency band, it becomes possible to implement filtering with a semiconductor circuit, namely, an active RC filter or the like in place of the SAW filter or the dielectric filter. 
     The channel filter suppresses signals present in a channel adjacent to the desired channel, a channel adjacent to the adjacent channel, and the like, that is, in the channels other than the desired channel. However, if a cutoff frequency which is a frequency having a gain of −3 dB from the DC gain of the channel filter shifts due to the manufacturing variation of the semiconductor, element temperatures, power supply voltage characteristics, etc., the received signal quality deteriorates. 
     For example, if the cutoff frequency shifts to the higher side, the degree of suppression of signals present in the adjacent channel, the channel adjacent to the adjacent channel, and the like deteriorates. If the cutoff frequency shifts to the lower side, the signal power of the desired channel decreases, so that the signal-to-noise ratio deteriorates, and the receiving sensitivity decreases. Further, in the case of receiving a digitally modulated signal, a deterioration in intersymbol interference characteristics affects the received data error rate. 
     Accordingly, a circuit for automatically adjusting the cutoff frequency of the channel filter is required, for example, as described in Patent Document 1. According to Patent Document 1, an automatic filter cutoff frequency adjusting circuit and a register for error correction are provided for each feedback capacitance and negative feedback capacitance (ground capacitance), thereby making it possible to adjust the filter cutoff frequency without increasing the error caused by the capacitance difference between the feedback capacitance and the negative feedback capacitance.
     [Patent Document 1] Japanese Unexamined Patent Publication No. 2009-94734   [Non-patent Document 1] The Design of CMOS Radio-Frequency Integrated Circuits, CAMBRIDGE, written by Thomas H. Lee   

     SUMMARY 
     As described above, according to Patent Document 1, an automatic filter cutoff frequency adjusting circuit and a register for error correction are provided for each feedback capacitance and negative feedback capacitance (ground capacitance), thereby making it possible to adjust the filter cutoff frequency without increasing the error caused by the capacitance difference between the feedback capacitance and the negative feedback capacitance. This makes it possible to automatically adjust the filter cutoff frequency of a large capacitance ratio to be used and reduce the time required for the automatic adjustment. 
     To form a filter that supports various cutoff frequencies, a digital capacitance which can be adjusted over a wider range is required. The present inventors have studied the automatic cutoff frequency adjusting technique of the filter having such a digital capacitance, and have found the necessity to automatically adjust the cutoff frequency of the filter to an arbitrary setting value within the adjustment range by further improving the cutoff frequency calibration accuracy. 
     It is an object of the present invention to provide an automatic cutoff frequency adjusting circuit which can automatically adjust the cutoff frequency of a filter to an arbitrary setting value within the adjustment range and a portable digital assistant having the automatic cutoff frequency adjusting circuit. 
     The above and other objects and novel features of the present invention will be apparent from the description of this specification and the accompanying drawings. 
     A typical aspect of the invention disclosed in the present application will be briefly described as follows. 
     An automatic cutoff frequency adjusting circuit includes a voltage/current converter circuit, a charge circuit, a discharge circuit, a digital capacitance having a plurality of electrostatic capacitances, a comparator for comparing a voltage inputted to the digital capacitance with a reference voltage, and a capacitance control circuit for controlling the digital capacitance. The digital capacitance is coupled to the charge circuit and the discharge circuit through a switch, a reset signal is inputted to the switch and the capacitance control circuit, all or some of the electrostatic capacitances of the digital capacitance are coupled in parallel by switching, the digital capacitance is coupled to the discharge circuit when the reset signal is at a first level, and the digital capacitance is coupled to the charge circuit when the reset signal is at a second level. The capacitance control circuit measures a time until the comparator detects that the voltage inputted to the digital capacitance is higher than the reference voltage after the reset signal has become the second level, and controls the digital capacitance by repeating, under a predetermined condition, processing for obtaining a next setting value of the digital capacitance, based on a measurement result, a target value of the digital capacitance, and the current value of the digital capacitance. 
     A typical effect of the invention disclosed in the present application will be briefly described as follows. 
     It is possible to provide the automatic cutoff frequency adjusting circuit which can automatically adjust the cutoff frequency of a filter to an arbitrary setting value within the adjustment range and the portable digital assistant having the automatic cutoff frequency adjusting circuit. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram showing a configuration example of an automatic cutoff frequency adjusting circuit according to a first embodiment of the present invention. 
         FIG. 2  is a flowchart of main processing in the automatic cutoff frequency adjusting circuit shown in  FIG. 1 . 
         FIG. 3  is a circuit diagram showing a configuration example of a digital capacitance included in the automatic cutoff frequency adjusting circuit shown in  FIG. 1 . 
         FIG. 4  is a table of assistance in explaining the relationship between digital values inputted to the digital capacitance and capacitance values between the input and output terminals of the digital capacitance. 
         FIG. 5  is a circuit diagram showing a configuration example of a channel filter circuit included in the automatic cutoff frequency adjusting circuit shown in  FIG. 1 . 
         FIG. 6  is a table of assistance in explaining the relationship among digital values and digital capacitance values of the digital capacitance and cutoff frequencies. 
         FIG. 7  is a flowchart of main processing in an automatic cutoff frequency adjusting circuit according to a second embodiment of the invention. 
         FIG. 8  is a block diagram showing a configuration example of a portable digital assistant incorporating the automatic cutoff frequency adjusting circuit according to the invention. 
     
    
    
     DETAILED DESCRIPTION 
     1. Summary of the Embodiments 
     Summary of a typical embodiment of the invention disclosed in the present application will first be explained. Reference numerals in the drawings that refer to with parentheses applied thereto in the description of the summary of the typical embodiments are merely illustration of ones contained in the concepts of components marked with the reference numerals. 
     [1] An automatic cutoff frequency adjusting circuit ( 412 ) according to a typical embodiment of the invention includes a voltage/current converter circuit ( 30 ), a charge circuit ( 50 ), a discharge circuit ( 40 ), a digital capacitance ( 70 ) having a plurality of electrostatic capacitances, a comparator ( 80 ) for comparing a voltage inputted to the digital capacitance with a reference voltage, and a capacitance control circuit ( 600 ) for controlling the digital capacitance. The digital capacitance is coupled to the charge circuit and the discharge circuit through a switch, a reset signal is inputted to the switch and the capacitance control circuit, all or some of the electrostatic capacitances of the digital capacitance are coupled in parallel by switching, the digital capacitance is coupled to the discharge circuit when the reset signal is at a first level, and the digital capacitance is coupled to the charge circuit when the reset signal is at a second level. The capacitance control circuit measures a time until the comparator detects that the voltage inputted to the digital capacitance is higher than the reference voltage after the reset signal has become the second level, and controls the digital capacitance by repeating, under a predetermined condition, processing for obtaining a next setting value of the digital capacitance, based on a measurement result, a target value of the digital capacitance, and the current value of the digital capacitance. 
     According to the above configuration, the time until the comparator detects that the voltage inputted to the digital capacitance is higher than the reference voltage after the reset signal has become the second level is measured, and the digital capacitance is controlled by repeating, under the predetermined condition, processing for obtaining the next setting value of the digital capacitance, based on the measurement result, the target value of the digital capacitance, and the current value of the digital capacitance. Thus, a cutoff frequency is automatically adjusted, and the digital capacitance is adjusted to the target value. The target value can be an arbitrary setting value within the adjustment range. 
     [2] In item [1], the capacitance control circuit includes a first counter (calcounter;  601 ), a second counter (counter;  602 ), a first register (target00;  603 ), and a second register (target0;  604 ). The time until the comparator detects that the voltage inputted to the digital capacitance is higher than the reference voltage after the reset signal has become the second level is measured using the second counter, the target value of the digital capacitance is set to the first register, the next setting value of the digital capacitance is stored in the second register, and the processing for obtaining the next setting value of the digital capacitance is repeated using the first counter. This makes it possible to easily achieve the function of the capacitance control circuit of item [1]. 
     [3] In item [2], the capacitance control circuit controls the digital capacitance so that a difference between a potential of the digital capacitance and the reference voltage falls within a predetermined potential difference, based on a comparison result by the comparator. 
     [4] In item [3], in the case of automatically adjusting the cutoff frequency of a channel filter circuit, the capacitance control circuit sets a digital capacitance within the channel filter circuit to the setting condition of the digital capacitance. Thereby, the cutoff frequency of the channel filter circuit is automatically adjusted. 
     [5] In item [4], to improve the calibration accuracy of the cutoff frequency, in the case where ΔC is the minimum capacitance step size of the digital capacitance and R 1  is a resistance for converting a DC voltage into a current by the voltage/current converter circuit, the capacitance control circuit sets the digital capacitance within the channel filter circuit in synchronization with a clock signal, and the frequency of the clock signal is set to be greater than 1/(ΔC·R 1 ). 
     [6] In item [4], the capacitance control circuit that supports a plurality of cutoff frequencies of the channel filter circuit has a plurality of target values as targets, a first digital capacitance setting value obtained by repeating the processing with a first target value corresponding to a first cutoff frequency is stored, and a second digital capacitance setting value obtained by repeating the processing with a second target value corresponding to a second cutoff frequency is stored. The capacitance control circuit uses the first digital capacitance setting value to set the first cutoff frequency, and the capacitance control circuit uses the second digital capacitance setting value to set the second cutoff frequency. This makes it possible to switch the cutoff frequencies at high speed. 
     [7] It is possible to incorporate the automatic cutoff frequency adjusting circuit of item [1] into a portable digital assistant. The portable digital assistant ( 801 ) incorporates the channel filter circuit ( 407 ), and the automatic cutoff frequency adjusting circuit automatically adjusts the cutoff frequency of the channel filter circuit. 
     2. Further Detailed Description of the Embodiments 
     Embodiments will be described in greater detail below. 
     First Embodiment 
       FIG. 8  shows a configuration example of a portable digital assistant according to the invention. 
     The portable digital assistant  801  includes an antenna  400 , a duplexer  401 , a low-noise amplifier (LNA)  402 , mixers  403 A and  403 B, a local oscillation circuit  404 , a 90-degree phase-shift circuit  405 , first amplifier circuits  406 A and  406 B, a channel filter circuit  407 , second amplifier circuits  408 A and  408 B, A/D (analog/digital) converter circuits  409 A and  409 B, a baseband LSI  410 , a transmission circuit  411 , and an automatic cutoff frequency adjusting circuit  412 . 
     The duplexer  401  passes only a desired band of a signal received from the antenna  400 . The LNA  402  amplifies the output signal of the duplexer  401  with low noise. The local oscillation circuit  404  oscillates at an integral multiple of a desired carrier frequency of the output signal of the duplexer  401 . The 90-degree phase-shift circuit  405  outputs an output signal I of the local oscillation circuit  404  and a signal Q obtained by shifting the phase of the signal I by 90 degrees. The mixers  403 A and  403 B mix the output signal of the LNA  402  with the output signals of the 90-degree phase-shift circuit  405 . The first amplifier circuits  406 A and  406 B amplify the output signals of the mixers  403 A and  403 B. The channel filter circuit  407  passes only a desired channel signal of the output signals of the first amplifier circuits  406 A and  406 B. The second amplifier circuits  408 A and  408 B amplify the output signals of the channel filter circuit  407 . The A/D converter circuits  409 A and  409 B convert the output signals of the second amplifier circuits  408 A and  408 B into digital signals. The baseband LSI demodulates the output signals of the A/D converter circuits  409 A and  409 B to receive the signals. The transmission circuit  411  transmits a signal. That is, the transmission circuit  411  amplifies a modulated output from the baseband LSI  410 , performs filtering and frequency conversion, and outputs the signal to the duplexer  401 . The duplexer  401  passes only a desired transmission band of the signal conveyed from the transmission circuit  411 . The signal that has passed through the duplexer  401  is transmitted from the antenna  400 . The cutoff frequency of the channel filter circuit  407  is automatically adjusted by the automatic cutoff frequency adjusting circuit  412 . 
       FIG. 1  shows a configuration example of the automatic cutoff frequency adjusting circuit  412 . 
     Although not limited in particular, the automatic cutoff frequency adjusting circuit  412  includes a resistor  20 , a voltage/current converter circuit (V/I)  30 , a discharge circuit (discharge)  40 , a charge circuit (charge)  50 , a switch  61 , a digital capacitance  70 , a comparator  80 , a latch circuit (Latch)  90 , the channel filter circuit  407 , and an arithmetic circuit  600 . 
     A reference voltage source generates a reference voltage  10 . The reference voltage  10  is a DC voltage VBG independent of process and temperature and power supply voltage. The value of the resistor  20  is R 1 . The voltage VBG is converted into a DC current I by the voltage/current converter circuit  30  and the resistor  20 . The DC current I is expressed by the following equation.
 
 I=VBG/R 1  (1)
 
     The DC current I is conveyed to the charge circuit  50 . The charge circuit  50  and the discharge circuit  40  are selectively coupled to the digital capacitance  70  by the switch  61 . The operation of the switch  61  is controlled by a reset signal  110 . Reference numeral  60  denotes a polarity diagram of the switch  61 . According to this, if the reset signal  110  is at an H level (corresponding to logical value “1”), the discharge circuit  40  is selectively coupled to the digital capacitance  70 , and if the reset signal  110  is at an L level (corresponding to logical value “0”), the charge circuit  50  is selectively coupled to the digital capacitance  70 . The digital capacitance  70  is set by the arithmetic circuit  600 . In this sense, the arithmetic circuit  600  is an example of a capacitance control circuit according to the invention. The channel filter circuit  407  includes a digital capacitance  151 . The digital capacitance  151  is set by the arithmetic circuit  600  so that the value of the digital capacitance  151  is equal to that of the digital capacitance  70 . The comparator  80  compares a level at the input terminal of the digital capacitance  70  with the reference voltage  10 . That is, the comparator  80  compares a voltage (VBG/R 1 /C 1 ×Δt) between the terminals of the digital capacitance  70  with the reference voltage  10  (VBG). The comparison result is held by the latch circuit  90 . The output of the latch circuit  90  is supplied to the arithmetic circuit  600 . The latch circuit  90  is initialized by the reset signal  110 . The arithmetic circuit  600  includes a first counter (calcounter)  601 , a second counter (counter)  602 , a first register (target00)  603 , and a second register (target0)  604 . 
     The digital capacitance  70  is configured as shown in  FIG. 3 . 
     The digital capacitance  70  shown in  FIG. 3  includes a plurality of electrostatic capacitances (simply referred to as capacitances) CDcom and CD 0  to CD 4  and switches SW 0  to SW 4 . The capacitances CD 0  to CD 4  are series-coupled to the respective switches SW 0  to SW 4 . The operations of the switches SW 0  to SW 4  are controlled by a digital value conveyed through a digital-value input terminal  320 . Reference numeral  300  denotes an input terminal, and reference numeral  310  denotes an output terminal. Further, reference numeral  330  denotes a polarity diagram of the switches SW 0  to SW 4 . In this example, if the digital value conveyed through the digital-value input terminal  320  is logical value “1”, the corresponding switch SW 0  to SW 4  is turned on. By the switching of the switches SW 0  to SW 4 , all or some of the capacitances CDcom and CD 0  to CD 4  are coupled in parallel. 
     A thick line shown in  FIG. 3  collectively represents five signal lines for five bits. It is possible to vary the capacitance value of the digital capacitance over a range of −32% to +30% in accordance with 5-bit digital values inputted through the digital-value input terminal  320 . 
     To achieve a variable range of −32% to +30%, since the capacitance CDcom is a capacitance value when all the switches SW 0  to SW 4  are turned off (when the L-level signal is inputted to all the switches SW 0  to SW 4  through the digital-value input terminal  320  according to the switch polarity diagram  330 ), the value of the capacitance CDcom is 0.68 C. In this context, “C” denotes the center of capacitance values to be achieved by the digital capacitance. To achieve a variable range of −32% to +30% with 5 bits, the digital capacitance is varied in steps of 2%; therefore, the value of the capacitance CD 4  is 0.32 C, the value of the capacitance CD 3  is 0.16 C, the value of the capacitance CD 2  is 0.08 C, and the value of the capacitance CD 1  is 0.04 C. 
       FIG. 4  shows the relationship between values provided to the switches SW 0  to SW 4  in accordance with 5-bit digital values inputted through the digital-value input terminal  320  and capacitance values between the input terminal  300  and the output terminal  310 . As shown in  FIG. 4 , the states of the switches SW 0  to SW 4  are controlled in accordance with the 5-bit digital values inputted through the digital-value input terminal  320 , thereby changing the value of the digital capacitance  70  over a variable range of −32% to +30%. 
       FIG. 5  shows a configuration example of the channel filter circuit  407  shown in  FIG. 1 . 
     The channel filter circuit  407  shown in  FIG. 5  is an eighth-order Butterworth low-pass filter configured with a positive feedback type low-pass circuit, and includes capacitances C 11  to C 42 , resistors R 11  to R 42 , and amplifiers AMP 1  to AMP 4 . Reference numeral  200  denotes an input terminal, and reference numeral  210  denotes an output terminal. The operational amplifiers AMP 1  to AMP 4  are used as voltage followers. The input terminal is coupled through the resistors R 11  and R 12  to the non-inverting input terminal (+) of the amplifier AMP&#39;. The output terminal of the amplifier AMP 1  is coupled through the resistors R 21  and R 22  to the non-inverting input terminal (+) of the amplifier AMP 2 . The output terminal of the amplifier AMP 2  is coupled through the resistors R 31  and R 32  to the non-inverting input terminal (+) of the amplifier AMP 3 . The output terminal of the amplifier AMP 3  is coupled through the resistors R 41  and R 42  to the non-inverting input terminal (+) of the amplifier AMP 4 . The output terminal  210  is coupled to the amplifier AMP 4 . The capacitance C 11  is coupled in parallel to the resistor R 12  and the amplifier AMP 1 . The capacitance C 21  is coupled in parallel to the resistor R 22  and the amplifier AMP 2 . The capacitance C 31  is coupled in parallel to the resistor R 32  and the amplifier AMP 3 . The capacitance C 41  is coupled in parallel to the resistor R 42  and the amplifier AMP 4 . The capacitance C 12  is coupled between the non-inverting input terminal (+) of the amplifier AMP 1  and the ground. The capacitance C 22  is coupled between the non-inverting input terminal (+) of the amplifier AMP 2  and the ground. The capacitance C 32  is coupled between the non-inverting input terminal (+) of the amplifier AMP 3  and the ground. The capacitance C 42  is coupled between the non-inverting input terminal (+) of the amplifier AMP 4  and the ground. 
     The higher frequency band of a signal inputted through the input terminal  200  than a desired cutoff frequency is suppressed by the low-pass filter including the amplifiers AMP 1  to AMP 4 , the capacitances C 11  to C 42 , and the resistors R 11  to R 42 , and the filtered signal is outputted through the output terminal  210 . The cutoff frequency of the channel filter circuit  407  varies in proportion to 1/(2π resistance variation×capacitance variation). 
     The capacitances C 11  to C 42  shown in  FIG. 5  are the digital capacitance  151 , and the capacitance value is changed by the digital value provided from the arithmetic circuit  600 , thereby making it possible to adjust the cutoff frequency of the filter. 
     Next, referring to a flowchart shown in  FIG. 2 , the operation of the above configuration will be described. 
     First, the reset signal  110  becomes the H level, so that the latch circuit  90  and the arithmetic circuit  600  are reset (S 201 ). Then, the first register (target00)  603  in the arithmetic circuit  600  is set to an initial value (target value) “target value”, the second register (target0)  604  in the arithmetic circuit  600  is set to the same value as the first register (target00)  603 , and the first counter (calcounter)  601  is set to N (N is an integer equal to or greater than 1) (S 202 ). 
     The value of the second register (target0)  604  is set to the switches SW 0  to SW 4  in the digital capacitance  70  (S 203 ). 
     Since the reset signal  110  is at the H level in step S 201 , the digital capacitance  70  is coupled to the discharge circuit  40  through the switch  61 , so that charge stored in the digital capacitance  70  is discharged by the discharge circuit  40 , and the voltage between the terminals of the digital capacitance  70  becomes 0 V. Thereby, the output of the comparator  80  becomes the L level. 
     Then, the value of the second counter (counter)  602  is set to 0 (S 204 ), and after a lapse of sufficient time for the discharge of the digital capacitance  70  (S 205 ), the reset signal  110  becomes the L level (S 206 ). 
     When the reset signal  110  becomes the L level, the digital capacitance  70  is coupled to the charge circuit  50  through the switch  61 , so that the DC current (I=VBG/R 1 ) which is the output of the voltage/current converter circuit  30  is supplied to the digital capacitance  70 . A voltage V proportional to a time Δt after the reset signal  110  has become the L level appears between the terminals of the digital capacitance  70 , as expressed by the following equation.
 
 V =( VBG/R 1/ C 1)·Δ t   (2)
 
     where C 1  denotes the capacitance value between the terminals of the digital capacitance  70 . 
     If the voltage V is larger than the voltage VBG, the output of the comparator  80  becomes the H level. If the voltage V is smaller than the voltage VBG, the output of the comparator  80  becomes the L level. 
     Every time a rising edge or falling edge of a clock signal  100  is observed until the output of the comparator  80  becomes the H level after the reset signal  110  has becomes the L level, the value of the second counter (counter)  602  in the arithmetic circuit  600  is incremented by 1 (S 207 , S 208 ). Therefore, the value of the second counter (counter)  602  in the arithmetic circuit  600  when the output of the comparator  80  has become the H level is equal to the total number of rising edges or falling edges of the clock signal  100  existing until the output of the comparator  80  becomes the H level after the reset signal  110  has become the L level. 
     A clock frequency Fc is expressed by the following equation in which ΔC is the minimum capacitance step size of the digital capacitance  70 . As shown below, sufficient accuracy can be obtained.
 
 Fc&gt; 1/(Δ C·R 1)  (3)
 
     The second counter (counter)  602  in the arithmetic circuit  600  is incremented until the output of the latch circuit  90  becomes the H level by the output of the comparator  80  (S 208 , S 209 ). When the output of the latch circuit  90  becomes the H level by the output of the comparator  80 , the second register (target0)  604  in the arithmetic circuit  600  is updated to a value expressed by the following equation.
 
target0=target00−counter+target0  (4)
 
     The new target0 value is set to the switches SW 0  to SW 4  in the digital capacitance  70  (S 210 ). This calibrates the manufacturing variation of the resistance value R 1  of the resistor  20  and the capacitance value C 1  between the terminals of the digital capacitance  70 , and the above digital value is provided to the switches SW 0  to SW 4  so as to make the product of R 1  and C 1  constant. 
     Then, the value of the first counter (calcounter)  601  in the arithmetic circuit  600  is decremented by 1 (S 211 ), and it is determined whether or not the value of the first counter (calcounter)  601  is 0 (S 212 ). If the value of the first counter (calcounter)  601  is not 0, the flow returns to step S 203  to repeat the steps after step S 203 . The steps after step S 203  are repeated until the value of the first counter (calcounter)  601  becomes 0. When the value of the first counter (calcounter)  601  becomes 0, a calibration end signal  160  is asserted. Thereby, the above digital value is set to the channel filter circuit  407  so as to calibrate the product of the resistance and the capacitance which determines the cutoff frequency of the filter circuit including the digital capacitance  151 . 
     Thus, according to the first embodiment, the digital value for calibrating and making the product of R 1  and C 1  constant is determined by the arithmetic circuit  600 , and the same digital value is provided to the capacitances C 11  to C 42  shown in  FIG. 5 , thereby setting the cutoff frequency of the channel filter circuit  407  to the target value. 
       FIG. 6  shows an example of cutoff frequencies obtained by applying digital values and capacitance values of the digital capacitance in  FIG. 6  to  FIG. 5  without the manufacturing variation of capacitance and resistance. 
     For example, in the case of setting the cutoff frequency to 5 MHz, the initial value “target value” in step S 202  in  FIG. 2  is set to a digital value 16, the digital value for calibrating and making the product of R 1  and C 1  constant is determined by the arithmetic circuit  600 , and the same digital value is provided to the capacitances C 11  to C 42  shown in  FIG. 5 , thereby automatically adjusting the cutoff frequency to a target frequency of 5 MHz. 
     Similarly, in the case of setting the cutoff frequency to 5.95 MHz, the initial value “target value” in step S 202  in  FIG. 2  is set to a digital value 8, the digital value for calibrating and making the product of R 1  and C 1  constant is determined by the arithmetic circuit  600 , and the same digital value is provided to the capacitances C 11  to C 42  shown in  FIG. 5 , thereby automatically adjusting the cutoff frequency to a target frequency of 5.95 MHz. 
     In the case of switching a plurality of cutoff frequencies, the digital value obtained by the arithmetic circuit  600  when the initial value “target value” is set to the digital value 16 is stored beforehand as CA in the register or the like, and the digital value obtained by the arithmetic circuit  600  when the initial value “target value” in step S 202  in  FIG. 2  is set to the digital value 8 is stored beforehand as CB in the register or the like. In the case of setting the cutoff frequency of the circuit shown in  FIG. 5  to 5 MHz, CA is provided to the digital value provided to the capacitances C 11  to C 42  shown in  FIG. 5 , and in the case of switching the cutoff frequency of the circuit shown in  FIG. 5  to 5.95 MHz, the digital value provided to the capacitances C 11  to C 42  shown in  FIG. 5  is switched to CB, thereby making it possible to switch the cutoff frequencies at high speed. 
     Second Embodiment 
     An automatic cutoff frequency adjusting circuit according to a second embodiment will be described. 
     The automatic cutoff frequency adjusting circuit according to the second embodiment differs from the circuit shown in  FIG. 1  mainly in how to raise the calibration end signal  160 . 
       FIG. 7  shows a flowchart of main processing in the automatic cutoff frequency adjusting circuit according to the second embodiment. The flowchart shown in  FIG. 7  differs from that shown in  FIG. 2  mainly in the addition of determination in step S 713 . Steps S 701  to S 712  in the flowchart shown in  FIG. 7  correspond to steps S 201  to S 212  in the flowchart shown in  FIG. 2 , and their detailed description is omitted. 
     After the new target0 value is set to the switches SW 0  to SW 4  in the digital capacitance  70  in step S 710 , it is determined whether or not the following equation holds, using preset +/− cal (calibration) threshold values, target0 and target00.
 
+cal threshold value≧(target0−target00)≧−cal threshold value  (5)
 
     The cal threshold values relate to the accuracy and speed of calibration. As the range of the +/− cal threshold values narrows, the accuracy increases, but the processing time increases. If equation (5) holds, the processing in this flowchart ends, the calibration end signal  160  is asserted, and the digital value is provided to the channel filter circuit  407 . If equation (5) does not hold, in step S 711  the value of the first counter (calcounter)  601  in the arithmetic circuit  600  is decremented by 1 (S 711 ). 
     Thus, if equation (5) holds in step S 713  using the cal threshold values, the processing in this flowchart ends, the calibration end signal  160  is asserted, and the digital value is provided to the channel filter circuit  407 , which can enhance the calibration speed. For example, in the case where there is no manufacturing variation, equation (6) holds; therefore, the arithmetic result of the arithmetic circuit  600  when the output of the comparator  80  has become the H level is expressed by equation (7). 
     
       
         
           
             
               
                 
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                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               00 
                             
                             + 
                             
                               target 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               00 
                             
                           
                           = 
                           
                             target 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             00 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
     Assume that the + cal threshold value=+1, and the − cal threshold value=−1. In comparison with the +/− cal threshold values, “target0−target00=0” satisfies the condition of equation (5); therefore, it is possible to complete the calibration by setting the capacitance one time in the second embodiment. In the first embodiment, it is necessary to set the capacitance N times, whereas in the second embodiment, it is possible to complete the calibration by setting the capacitance one time, which enables the high-speed calibration. 
     Further, if “target0−target00” does not satisfy the condition of equation (5), with the first counter (calcounter)  601  it is possible to complete the calibration without fail by setting the capacitance N times as in the first embodiment. 
     While the invention made above by the present inventors has been described specifically based on the illustrated embodiments, the present invention is not limited thereto. It is needless to say that various changes and modifications can be made thereto without departing from the spirit and scope of the invention. 
     For example, the channel filter circuit  407  shown in  FIG. 5  is not limited to the Butterworth filter. Even if the channel filter circuit  407  is a filter having arbitrary characteristics such as a Chebyshev filter, the automatic filter cutoff frequency adjusting circuit  412  shown in  FIG. 1  can automatically adjust the cutoff frequency. Further, the channel filter circuit  407  shown in  FIG. 5  is not limited to the positive feedback type low-pass circuit. Even if the channel filter circuit  407  has a different configuration such as a biquad filter, the automatic filter cutoff frequency adjusting circuit shown in  FIG. 1  can automatically adjust the cutoff frequency. 
     Further, the reference voltage  10  generated by the reference voltage source is the DC voltage VBG independent of temperature and power supply voltage, but is not limited thereto. For example, the reference voltage  10  may be generated by dividing the power supply voltage through the use of resistors, which can generate the reference voltage  10  independent of temperature and power supply voltage in an environment with little fluctuation in power supply voltage.