Patent Publication Number: US-8525442-B2

Title: Operating circuit for LEDs

Description:
The invention relates to an operating circuit provided with light diodes according to the preamble of the patent claim  1 , and to a method for operating light diodes according to the preamble of patent claim  14 . 
     TECHNICAL FIELD 
     Semiconductor light sources, such as for example light diodes, have been increasingly attracting attention in recent years in particular with regard to their application for illumination. One of the reasons for this is the fact that important technical innovations and major advances have been achieved both with respect to brightness, but also with respect to light efficiency (the light output per Watt) of these light sources. Las but not least, it also became possible to develop light diodes which thanks to their relatively long lifespan represent an attractive alternative to conventional light sources such as glow (incandescent) lamps or gas discharge lamps. 
     PRIOR ART 
     Semiconductor light sources are well known from prior art. Hereinafter, they will be abbreviated as LEDs (light emitting diodes). This term will include in the following text both light diodes that are made from inorganic materials, as well light diodes that are made from organic materials. It is known that the light irradiation from LEDs is correlated with the current flowing through the LEDs. 
     In order to control brightness, LEDs are essentially operated in a mode in which the flow of the current through the LED is regulated. 
     A switch controller (step-down or buck controller) is preferably used in practice in order to control an arrangement containing one or more LED. A similar switch controller is known for example for DE 10 2006 034 371 A1. In this case, a control unit controls a switch which is clocked at a high frequency (for example a power transistor). In the activated stat of the switch, the current flows through the LED arrangement and a coil which is charged in this manner. The energy of the coil, which is stored with intermediate storage, is discharged through the LEDs (recovery phase). The current displays a zigzag path through the LEDs: when the switch is in the activated state, the LED current displays a climbing edge, when the switch is turned off, a trailing edge is displayed. The mean value of the time interval of the LED current represents the effective current flowing through the LED arrangement as a measurement of the brightness of the LEDs. The mean effective current can thus be controlled with a suitable clocking of the power switch. 
     The function of the operating device is in this case to adjust the current flowing through the LED to a desired mean current flow, and to maintain the temporal fluctuations in the range of the variations of the current, which will depend on the high frequency that is used to turn the switch on and off (typically in the range above 10 kHz), at a level that is as low as possible. 
     A wide range of variations of the current (waviness or ripples) exerts a particularly detrimental influence on the LEDs because the spectrum of the emitted light can be changed when the amplitude of the current is changed. 
     In order to maintain the spectrum of the emitted light as constant as possible, it is known that instead of varying the amplitude of the current to control the brightness of LEDs, a so called PWM (pulse width modulation) method can be used. The LEDs are in this case maintained supplied using the lower frequency pulse packets of the operating device (typically with a frequency in the range from 100-1,000 Hz) at a constant current amplitude (on a time average). The current in one pulse packet is superimposed on the high frequency ripple mentioned above. The brightness of the LEDs can be then controlled by the frequency of the pulse packet, the LEDs can be for example dimmed so that the time interval between the pulse packets is increased. 
     A practical requirement on the operating device is that it should be possible to use the device universally and with as much flexibility as possible, for example, independently of how many LEDs representing a load are in fact connected and operated. Moreover, it should be possible to change the load during the operation, for example when one LED fails. According to conventional technology, the LEDs are operated in a so called “continuous conduction mode”, or with continuous operations. This method will be explained in greater detail with reference to  FIG. 1   a  and  FIG. 1   b  (prior art). 
     In the example indicated in  FIG. 1   a , a step-down converter (buck converter) serves as a basic circuit for the operation of one LED (or several LEDs connected in series), which is equipped with a first switch S 1 . Direct current voltage or rectified alternating current voltage U 0  is supplied to the operating circuit. 
     When the first circuit S 1  is turned on, energy is built up in the coil L 1  (during the time period t_on), which is then discharged in the turned off state of the first switch S 1  (time period t_off) through at least one LED. The resulting current profile with respect to time is illustrated in  FIG. 1   b  (conventional technology). Two pulse packets PWM are indicated in this case. The current profile within one pulse packet is shown at a magnified scale. In order to maintain a constant color, the amplitude of the ripple within the pulse packet should be as small as possible. This can be achieved with a suitable selection of the point in time t 0  when the device is turned on, and the point in time t 1 , which is the point when the device is turned off. These points in time can be selected for example so that the first switch S 1  is activated when the current is below a certain minimum reference value, and so that the switch is turned off when the maximum reference value is exceeded. This method has several disadvantages: For one, a quick sequence of activating and deactivating occurrences is required to achieve a ripple that is as small as possible. This is because the increase (positive or negative edge) of the current cannot be controlled by the operating device, which should thus be considered as a given, since it is determined, among other things, by the inductance of the coil L 1  and by the increased output of the LEDs. 
     In order to reduce the waviness (ripples), more switching occurrences would have to take place within a certain time segment, which would naturally lead to switching losses. Moreover, these switching losses are particularly high in the continuous conduction mode. 
     Although an actual semiconductor switch is switched on very quickly, it cannot be switched on with infinite quickness. The amount of energy which is dissipated during the switching process will be greater when the switching process takes longer, and also when the output, which is applied during the switching process to the switch, is higher. In non-continuous operations, the switching losses are particularly high because the switching processes are realized when high currents are applied. 
     DESCRIPTION OF THE INVENTION 
     The object of the present invention to provide an operating circuit which is an improvement of prior art, for at least one LED, and a method for enabling the operation of at least one LED, which makes it possible to maintain in a simple manner a constant current and thus also the LED performance. 
     This object is achieved in accordance with the invention based on the independent claims of the invention. The dependent claims represent a further development of the central concept of the invention in a particularly advantageous manner. 
     According to a first aspect of the invention, the operating circuit for at least one LED supplies direct current voltage or rectified alternating current voltage. A supply voltage for at least one LED is provided by means of one coil and a first switch which is clocked by a control/regulation unit, whereby when the first switch is turned on, power is temporarily stored in the coil and it is discharged through a diode and through at least one LED when the first switch is turned off. 
     The operating circuit is equipped with a capacitor, which is arranged in parallel to at least one LED and which maintains the current during the phase of the demagnetization of the coil L 1  through the LED, so that the current through the LEDs is smoothed. 
     In accordance with the circuit according to the invention, the control/regulation unit selects the point in time for turning the first circuit off in such a way that as little switching losses as possible will occur, and despite the current flow provided through the at least one LED, a ripple that is as small as possible is exhibited. 
     In a preferred embodiment of the invention, the operating unit is equipped with a first sensor unit which generates a sensor signal with the first switch depending on the current flow, and/or a with second sensor unit which detects the demagnetization of the coil and generates a sensor signal. The sensor signals are supplied to and are processed by the control/regulation unit. 
     In accordance with the invention, the control unit uses a signal of the first sensor unit, or a signal of the second sensor unit, or a combination of both signals in order to determine the point in time for turning on and/off the first switch. 
     In accordance with the invention, the control/regulation unit turns the first switch off when the current flowing through the first switch exceeds a maximum reference value, and then it is turned on again at the point in time when the coil is demagnetized and/or the diode is blocking. In a preferred embodiment of the invention, the first sensor unit is a measurement resistor (shunt). In another preferred embodiment of the invention, the second sensor unit is a secondary winding which is inductively coupled with the coil, or a Hall sensor. In another embodiment, the second sensor unit senses that the demagnetization of the coil has been reached as it monitors the voltage above the first switch by means of a (resistive) voltage distributor. 
     Other preferred embodiments and further developments of the invention are the subject of further dependent claims. 
    
    
     
       The present invention will now be explained in detail based on preferred embodiments with reference to the attached drawings. 
         FIG. 1   a  shows a circuit arrangement according to the known prior art. 
         FIG. 1   b  shows a diagram indicating the progress in time of the LED current in the circuit arrangement of  FIG. 1   a  (prior art), 
         FIG. 2   a  shows a first example of an operating circuit (Buck) for LEDs according to the invention, 
         FIG. 2   b  shows a diagram, which illustrates current developments depending on time and control signals of the circuit arrangement shown in  FIG. 2   a,    
         FIG. 3  and  FIG. 4  show particular embodiments of the invention, 
         FIG. 5  shows a modification of the circuit of  FIG. 2   a  (Buck Boost), 
         FIG. 6  shows another particular embodiment of the invention. 
     
    
    
       FIG. 1   a  and  FIG. 1   b  show prior art. 
     The circuit arrangement shown in  FIG. 2   a  is used to operate at least one LED (or several LEDs, which are connected in series and/or in parallel). In the example shown in the drawing, two LEDs are used connected in series, but it is naturally also possible, for example, to use one or multiple LEDs. The LED or the LEDs that are connected in series or in parallel will be hereinafter referred to as LED segment. One advantage of the present invention is that the operating circuit can be adjusted with a great deal of flexibility depending on the type and number of the LEDs that are connected in series. The direct current voltage supplied to the circuit can naturally also be a rectified alternating current voltage. The LEDs are connected in series with a coil L 1  and with a first switch S 1 . 
     In addition, the circuit arrangement is equipped with a diode D 1  (the diode D 1  is connected to the LEDs and to the coil L 1  in parallel), and to a capacitor C 1  which is connected to the LEDs in parallel. In the activated state of the first switch S 1 , the current flows through the LEDs and through the coil L 1  which will become magnetized as a result. 
     In the turned off state of the first switch S 1 , the energy stored in the magnetic field of the coil is discharged in the form of a current flowing through the diode D 1  and the LEDs. In parallel with these occurrences, the capacitor C 1  is charged at the beginning of the activation of the first switch S 1 . 
     During the switched off phase of the first switch S 1  (recovery phase), the capacitor C 1  is discharged and it contributes to the current flowing through the LED segment. This can lead to a smoothing of the current through the LEDs with a suitable dimensioning of the capacitor C 1 . 
     For the first switch S 1 , a field effect transistor is preferably used, but a bipolar transistor can be also employed. The first switch S 1  is connected with a high frequency, typically in a frequency range above 10 kHz. 
     One advantage of the invention is that the first switch S 1  can be protected during operations because, as will be explained later, it is preferably activated when the power applied to it equals almost zero. In contrast to that, since switching events according to conventional technology occur at a high power, an expensive structural element must be used for the first switch S 1  with a very short duty cycle in order to maintain the switching losses within a tolerable range. 
     One advantage of the switch according to the invention is that for the first switch S 1  and for the diode D 1  a relatively much less expensive structural element can also be used with a relatively somewhat longer switching duration or longer depletion time. 
     The circuit of  FIG. 2   a  is further also provided with a control and/or regulation unit SR which provides the clock of the first switch S 1  used to control the LED power. 
     The control/regulation unit SR uses signals from a first sensor unit SE 1  and/or signals from a second sensor unit SE  2  as input size signals for determination of the exact point in time for turning the first switch S 1  on and off. 
     The first sensor unit SE 1  is arranged in series with the first switch S 1  and detects the current flowing through the first switch S 1 . This is used to monitor the current flowing through the first switch S 1 . If the current flowing through the first switch S 1  exceeds a certain predetermined maximum reference value, the first switch S 1  is turned off. In an advantageous embodiment of the invention, the first sensor unit SE 1  can be for instance a measurement resistor (shunt or current resistor). 
     In order to monitor the current flow, the voltage drop can be measured at the measurement resistor (shunt), for example by means of a comparator which performs a comparison to a reference value. 
     If the voltage drop at the measurement resistor (shunt) exceeds a predetermined value, the first switch S 1  is turned off. 
     The second unit SE 2  is arranged within the current branch through which the current flows during the recovery phase, preferably in the vicinity of or on the coil L 1 . By means of the second sensor unit SE 2 , the control unit/regulation unit SR can determine a suitable point in time for turning on the first switch S 1 . 
     In accordance with the invention, the first switch S 1  is preferably activated when the current flowing through the coil L 1  is for the first time at zero or at least very low, which is to say preferably during the time interval when the diode D 1  is blocking at the end of the recovery phase. According to the invention, as little current as possible is applied at the switch S 1  at the point in time when the switch S 1  is turned on. An almost loss-free switch is made possible with the detection of the zero crossing of the current at the coil L 1 . According to the invention, the current is indicated through the LEDs with only a small brightness and the fluctuations are not strong. This is due to the smoothing effect of the capacitor C 1  which is connected in parallel to the LEDs. The capacitor C 1  takes over the supplying of the LED during the phase of a low coil current. 
     The individual current characteristics and the optimal point in time for turning on the first switch S 11  will be further explained based on the diagram shown in  FIG. 2   b.    
     Similarly to the diagram shown in  FIG. 1   b , the figure shows the temporal course of the current i_L across two pulse packets. 
     The enlarged representation shows the current characteristics within a PWM pulse packet: namely the course in time of the current i_L in the coil L 1 , the course in time of the current i_LED in the LEDs, and the course in time of the state of the first switch S 1  (the first switch S 1  is turned off in state  0 , and the first switch S 1  is closed in state  1 ; the signals for the state of the switch S 1  correspond to the control signal (namely at the gate) of the switch S 1 . At the point in time t_ 0 , the first switch S 1  is closed and a current is starting to flow through the LED and through the coil L 1 . The current i_L displays an increase according to an exponential function, whereby a more or less linear increase of the current i_L can be recognized in the region which is of interest here. i_LED differs from i_L in that one part of the current i_L contributes to the charging of the capacitor C 1 . A consequence of the opening of the first switch S 1  at the point in time t_ 1  (for example when a desired maximum reference value has been reached) is that energy that is stored in the magnetic field of the coil is discharged through the diode D 1  and the LEDs or the capacitor C 1 . The current i_L flows then in the same direction, but it is continuously decreased and it can even reach a negative value. A negative current (which means a current in the opposite direction) will be present as long as the depletion of the charging carrier, which was previously enriched in the conducting polarized diode, is performed from the blocking layer of the diode D 1 . 
     The current i_LED will be increased only slightly and it will be maintained because the capacitor C 1  has a smoothing effect. At the point in time t_ 2 , the diode will be blocking. The current i_L is decreased (and will then be negative), and continues toward zero. In this phase, parasitic capacities on the diode D 1  and other parasitic capacities are transferred to the remaining circuit. 
     The voltage at the node Ux above the first switch S 1  drops to a low value (as a result of the blocking by the diode D 1 ). An advantageous point in time t_ 3  for reactivating the first switch S 1  will then be encountered when the current i_L reaches the zero crossing i_L, or at least when it is in the vicinity of the zero crossing. At this point in time, the coil L 1  is not magnetized or hardly magnetized at all. The first switch S 1  can be turned on at this point with very little loss because hardly any current is flowing through the coil L 1 . 
     However, it can be then reactivated again already at the point in time t_ 2 , or shortly prior to this point, because the current flowing through the coil L 1  is during this time period very low. 
     A second sensor unit SE 1  is used for detection of an advantageous point in time for turning on the first switch S 1 . In a first embodiment, the current i_L can be detected for example through the coil L 1 . However, this will necessitate relatively expensive circuits. The current i_L flowing through the coil L 1  can be detected for example by means of a Hall sensor. Additionally or as an alternative, further/other sizes can be utilized which are suitable for detection of an advantageous point in time for activation. 
     In another advantageous embodiment, for example the magnetization state of the coil L 1  can be detected. The second sensor unit SE 2  can be for example a secondary winding L 2  on the coil L 1 , which detects the voltage on the coil L 1 . The monitoring of the temporal progress of the voltage on the coil L 1 , (in particular of the “surge” shortly after the closing of the diode D 1  after the point in time t_ 2 ), makes it possible to obtain a determination of an advantageous point in time for the reactivation of the first switch S 1 . In a simple embodiment variant, a comparator would serve to detect that demagnetization (and thus the zero crossing) was reached based on a value above or below the threshold value. 
     Instead of or complementary to monitoring of the voltage on the coil L 1 , the voltage can also be monitored at the node Ux above the first switch S 1 . The voltage at the node Ux falls significantly from a high value to a low value when the diode is closed. The signal to reactivating the first switch S 1  can therefore be triggered when the voltage Ux is below a certain threshold value. The control/regulation unit SR turns the first switch S 1  on again at the point in time when the coil L 1  is demagnetized and/or the diode D 1  is blocking. The second sensor unit S 2  can in this cases comprise a secondary coil L 2  which is inductively coupled to the coil L 1 , or a voltage distributor (R 1 , R 2 ) at the node Ux. 
     The control/regulation unit SR uses the information obtained from the first sensor unit SE 1  and/or the second sensor unit SE 2  in order to determine the point in time for turning the first switch S 1  on and off. The regulation of the (time-averaged) LED performance through the control/regulation unit SR can be realized for example in the form of PWM signals. The frequency of the PWM signals is typically in the range of 100-1,000 Hz. 
       FIG. 3  and  FIG. 4  show particular embodiments of the invention. 
       FIG. 3  illustrates a particular embodiment of the circuit arrangement explained above (a step-down or buck converter). The advantageous point in time for deactivation is in this case obtained by detecting the voltage at the node Ux above the first switch S 1 . This is performed with the resistive distributors R 1  and R 2 . The node Ux is located between the coil L 1 , the diode D 1  and the switch S 1 . 
     A capacitive voltage distributor or a combined voltage distributor from resistance and capacitance can be used, for example, as a voltage distributor. 
     The measurement resistor (shunt) Rs serves to detect the current flowing through the first switch S 1 . The monitoring of the temporal progress of the voltage at the node Ux (in particular of the “drop” shortly after the blocking of the diode D 1  in the vicinity of the point in time t_ 2 ) enables a determination of the advantageous point in time for reactivating the first switch S 1 . 
     Instead of or complementary to voltage monitoring, the voltage can be monitored on the coil L 1  for example at the node Ux above the first switch S 1 . The voltage at the node falls significantly when the diode is blocking, from a high value to a low value. The signal to turn the first switch on again can therefore be triggered when the voltage of Ux is below a certain threshold. 
     In the switching arrangement of  FIG. 3 , a second switch S 2  is additionally arranged also parallel to the capacitor C 1 . The second switch S 2  can be controlled selectively/independently, and it can be for example a transistor (a MOSFET or a bipolar transistor). When the second switch S 2  is closed, the discharging operation of the capacitor S 2  is accelerated. The purpose of accelerated discharging of the capacitor C 1  is for the current flowing through the LED to attain the zero crossing point as soon as possible. This is desirable for example at the end of a PWM packet where the current flowing through the LED should fall off as quickly as possible, i.e. the falling edge of the current progress should be as abrupt as possible (to achieve constant colors). Preferably, the second switch S 2  can be activated and controlled at a low dimming level so that the PWM packets are very short, and it is important that the current flow through the LED at the end of a pulse packet quickly approach zero. For example, an even lower dimming level can be achieved with a suitable control of the second switch S 2 . 
     A further function of this second switch S 2  is that it bridges over in the activated state the LEDs. This is required for example when the LEDs are to be turned off, i.e. when they should not be emitting any light, but the U 0  supply is still applied. Without the bridging over through the second switch S 2 , a current (albeit a small one) would flow through the LEDs and the resistors R 1  and R 2  and the LED would be (weakly) lit up. 
     It should be mentioned that the arrangement provided with a second switch S 2  parallel to the LEDs and to the capacitor C 1  in order to accelerate the discharging of the capacitor C 1 , or to bridge the LEDs, is not limited only to special embodiments of the circuit arrangement of  FIG. 3 , since it can be employed in all embodiments of the invention. 
       FIG. 4  shows a modification of the circuit shown in  FIG. 3 , wherein the monitoring of the voltage is performed on the coil L 1 . The voltage on the coil L 1  can be detected for example with a secondary winding L 2  which is coupled to the coil L 1  (or an additional coil L 2  which is inductively coupled to the coil L 1 ). A secondary winding L 2  is used in this case in order to detect an advantageous point in time for the first switch S 1 . The monitoring of the temporal progress of the voltage on the coil L 1 , (in particular the “drop” in the vicinity of the closing of the diode D 1  after the point in time t_ 2 ), makes it possible to obtain a determination of an advantageous reactivation point for the first switch S 1 . This monitoring can, as was already mentioned, be performed also with a secondary coil L 2 . 
     The determination of the point in time for the zero crossing or for the demagnetization can also be achieved, as was already mentioned, by monitoring of the threshold values (based on monitoring of values below and above a threshold value, so that when the monitoring is performed by means of a secondary winding L 2 , the polarity of the voltage depends on the winding direction of the secondary winding L 2  relative to the coil L 1 ). 
     It should be mentioned that the method for detecting an advantageous point in time for turning on the first switch S 1  can be naturally also applied to a different switching topologies, for example for a so called flyback converter, or buck boost converter or forward converter.  FIG. 5  shows a modification of the circuit of  FIG. 2   a , in that the arrangement of the choke L 1 , of the diode D 1  and the orientation of the LED segment are modified (a flyback converter or a buck boost converter is formed). 
     A further development of the invention is illustrated in  FIG. 6 . The identification of the demagnetization of the coil L 1  can be carried out by means of monitoring of the voltage at the coil L 2 , for example with a standard control circuit IC which is readily available. This control circuit IC (integrated switching circuit), which corresponds to or contains the control/regulation unit SR according to  FIG. 2 through 5 , is equipped with an input for identifying that demagnetization of one coil was reached based on the monitoring of the voltage on one secondary winding provided on one of the coils. Further, the control circuit IC is equipped with an output for controlling one switch and with further monitoring inputs. 
     A first input of these monitoring inputs can be used for the input of a reference value such as for example reference voltage. 
     A second monitoring input can be used for monitoring whether a maximum voltage was reached, or also whether a maximum voltage is used based on measurement of voltage. A third monitoring input can be used to monitor another voltage, or also for the deactivation or activation of the control circuit IC, or to control the switches which are controlled by the control circuit IC. 
     As shown in  FIG. 6 , the control circuit monitors the current with a first switch S 1  during the phase when the first switch S 1  is turned on through the measurement resistors (shunt) Rs and through the input  4  at the control circuit IC. As long as the voltage, which is detected through measurement resistors (shunt) Rs, is detected, a certain maximum value is reached and the first switch S 1  is opened. The default value for a voltage level required for the opening of the first switch S 1  can be adjusted according to the default reference value at the input  3  of the control circuit IC. For example, a reference voltage can be input from a microcontroller, which inputs the maximum level through the measurement resistor (shunt) Rs of allowable voltage and thus also the maximum allowable current through the first switch S 1 . 
     By way of an example, the microcomputer can generate a PWM signal which is smoothed by a filter  10  (for example an RC element), and applied in this manner as a direct current voltage signal with a certain amplitude at the input  3  of the control circuit IC. The amplitude of the signal at the input  3  of the control circuit IC can be adjusted by changing the detecting relationships of the PWM signal of the microcontroller. 
     The control circuit IC can detect whether demagnetization of the coil L 1  has been reached via the input  5  based on the monitoring of the voltage on one of the secondary winding L 2 . This recognition can be used as a reactivation signal. As soon as the demagnetization of the coil L 1  has been recognized by the control circuit IC, the control circuit IC can switch on the first switch S 1  through the control applied via the output  7 . 
     The control circuit IC can perform activation and/or also deactivation by applying a voltage at the input  1 . This voltage for activation at the input  1  can be also alternated between a high and a low level, in which the control circuit IC is activated at the high level and at least the control over the first switch S 1  is performed at the low level. This control over the input  1  can be carried out by a microcontroller. For example, a low frequency activation and deactivation of the control circuit IC can be achieved in this manner and thus also the control over the first switch S 1 , as well as a low-frequency control over the operating circuit which can be used to dim the LEDs. 
     Moreover, it is also possible to set also another reference voltage for the control circuit IC through the amplitude of the signal that is applied at this input. This voltage can for example influence also the level of the maximum allowable current flowing through the switch, but also the allowable duration of the time period when the switch S 1  is turned on. The control circuit IC and/or the control circuit IC combined with the microcontroller can together form the control unit SR. 
     The duration of the activation period of the first switch S 1  can be also dependent on another voltage measurement within the operating circuit. For example, a Vsense voltage can be also supplied to the control circuit IC. It is thus also possible to perform measurements of the voltage at the node between the coil L 1  and the LED through a voltage distributor R 40 /R 47 . This voltage measurement Vsense can be supplied either to another input of the control circuit IC as an additional variable which is added to an already occupied input of the control circuit IC, but it can be also furnished to an input of the microcontroller. 
     Therefore, a system can be built in this manner which on the one hand enables a simple control of the dimming of the LED through a low-frequency PWM, and which on the other hand also makes it possible to achieve high-frequency operation of the operating device with as low losses as possible, in combination with as constant a current supplied through the LED as possible. 
     It is also possible to specify through a microcontroller both the frequency and the duty cycle of a PWM signal in order to dim the LEDs, and also the level of the maximum allowable current can be preset through the first switch S 1 . The microcontroller can control with a signal which is supplied to the input  1  of the control circuit IC the dimming of the LED using a low frequency PWM. In addition, the microcontroller can also specify through a signal which is supplied to the input  3  of the control circuit IC the level of the maximum current through the first switch S 1 , but also the required time period during which the first switch S 1  is turned on. 
     The operating circuit can further also contain another switch S 2  which is arranged in such a way that this second switch S 2  can bridge the LEDs. 
     The second switch S 2  can be further arranged in such a way that it can take over from the LEDs, or interrupt the current with an existing high resistive voltage measurement path, or a similar existing highly resistive switching arrangement. 
     The LEDs can be bridged and thus deactivated with a parallel switch of the second switch S 2  to the LEDs. This method can be used to adjust the brightness (dimming) of the LEDs. One possible variant would be when the dimming is realized via a second switch S 2 , while the current is adjusted and controlled only by controlling the first switch S 1 . 
     It is also possible to combine the control over both switches S 1  and S 2  to achieve an optimized dimming control. For example, the second switch S 2  can be additionally used only for dimming to a low dimming level. The operating switch is due to the existing topology and the control circuit is designed in such a way that the output voltage of the operating circuit (i.e. the voltage for the LEDs) is limited to a maximum allowable value. If the LEDs are bridged when the second switch S 2  is turned on, the control circuit limits the output voltage in such a way that no excessive current can flow which could result in possible interference. This control of the second switch S 2  can be used for example only for dimming on a low dimming level. 
     When the step down converter (buck converter) design is operated during the power source supplying operations (in a so called hysteretic mode such as the one described in the examples of the embodiments), the LEDs can be dimmed only with a second switch S 2  which has a very low impedance, the losses are very low. 
     In addition, the second switch S 2  can be controlled so that it can control the current with an available highly resistive voltage measurement path or a similar available highly resistive circuit arrangement taking charge of the LED. 
     When the first switch is not clocked, for example as shown in  FIG. 6 , no current should be flowing through the LEDs. However, because of the present voltage distributor R 40 /R 47 , a very small current can still flow through the LED. 
     In this case, the second switch S 2  can be closed with the desired deactivation of the LEDs (for example when no light should be emitted), so that the current flowing through the LEDs is interrupted or prevented from flowing. 
     The second switch S 2  can be controlled at least always in connection with a low frequency PWM packet in order to bridge or deactivate the LEDs (during the last discharge edge, which is to say at the end of a PWM pulse packet). 
     An interruption of the current flowing through the LEDs can be realized with an arrangement of the second switch S 2  in series with the LEDs. 
     The example of  FIG. 6  (and naturally as well as the other examples) can obviously also be further developed when several operating circuits according to  FIG. 6  are available. The control circuits IC or control units SR of the individual operating circuits are controlled by a common microcontroller. The individual operating circuits can be controlled for example by LED strings which have different wavelengths or colors. The control of the microcontroller can be realized via an interface (wireless or with fixed connections). The control signals can be in this case transmitted in order to adjust the brightness or color, or also status information, through the interface.