Patent Publication Number: US-2004057503-A1

Title: Method and apparatus for processing a composite signal including a desired spread spectrum signal and an undesired narrower-band interfering signal

Description:
FIELD OF THE INVENTION  
       [0001] This invention relates in general to communication systems, and more specifically to a method and apparatus for digitally processing a composite signal such as a desired spread spectrum signal and an undesired narrower-band interfering signal.  
       BACKGROUND OF THE INVENTION  
       [0002] In spread spectrum communications, the signals of interest are spread over a wide frequency range. This may result in superior noise immunity, improved system capacity, and improved signal integrity at the receive antenna. Unfortunately, wide-band signaling also increases the likelihood that the wireless communications physical channel will encounter spurious narrow-band interfering signals. This can occur as a result of either over-the-air interference or nonlinearities associated with analog components used in the end-to-end communications system.  
       [0003] In modern digital communications systems, the analog-to-digital converter is moving ever closer to the antenna to enable the use of Digital Signal Processing (DSP) techniques as close as possible to the signal capture point in the receiver. This has lead to a desire to implement entirely digital cancellation techniques for narrow-band interference in spread spectrum signals. Unfortunately, the high sampling rate associated with spread spectrum signals has implied high rate multiplication operations, specialized multiplication hardware, and extraordinarily high computational requirements for prior-art adaptive nulling techniques, such as the Least Means square (LMS) and the Recursive Least Squares (RLS) techniques. Furthermore, there has been an added complexity for properly tracking a time-varying frequency spur.  
       [0004] Thus, what is needed is a method and apparatus for digitally processing a composite signal such as a desired spread spectrum signal and an undesired narrow-band interfering signal. Preferably the processing will not require high rate multiplication operations.  
     
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
     [0005] The accompanying figures, where like reference numerals refer to identical or functionally similar elements throughout the separate views and which together with the detailed description below are incorporated in and form part of the specification, serve to further illustrate various embodiments and to explain various principles and advantages all in accordance with the present invention.  
     [0006]FIG. 1 is an electrical block diagram of an exemplary first embodiment of an apparatus.  
     [0007]FIG. 2 is a frequency-domain diagram depicting the magnitude over frequency of an exemplary composite signal.  
     [0008]FIG. 3 is an electrical block diagram of an exemplary second embodiment of an apparatus.  
     [0009]FIG. 4 is an electrical block diagram of an exemplary third embodiment of an apparatus.  
     [0010]FIG. 5 is a frequency-domain diagram depicting the magnitude over frequency of an exemplary shifted composite signal.  
     [0011]FIG. 6 is a frequency-domain diagram depicting the magnitude over frequency of an exemplary interference-free shifted signal.  
     [0012]FIG. 7 is a frequency-domain diagram depicting the magnitude over frequency of an exemplary recovered spread spectrum signal.  
     [0013]FIG. 8 is a communication device at least partially in block diagram form.  
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
     [0014] In overview form the present disclosure concerns communications systems that utilize receivers to provide service for communications units or more specifically a user thereof operating therein. More particularly, various inventive concepts and principles embodied as methods and apparatus for processing a composite signal including, for example, a desired spread spectrum signal and an undesired narrow-band interfering signal for use in equipment with such communications systems will be discussed and disclosed. The communications systems of particular interest are those being deployed and developed such as CDMA (Code Division Multiple Access), W-CDMA (Wideband-CDMA), CDMA2000, 2.5G (Generation), 3G, UMTS (Universal Mobile Telecommunications Services) systems and evolutions thereof that utilize spread spectrum signals, although the concepts and principles have application in other systems and devices, such as modems.  
     [0015] The instant disclosure is provided to further explain in an enabling fashion the best modes of making and using various embodiments in accordance with the present invention. The disclosure is further offered to enhance an understanding and appreciation for the inventive principles and advantages thereof, rather than to limit in any manner the invention. The invention is defined solely by the appended claims including any amendments made during the pendency of this application and all equivalents of those claims as issued.  
     [0016] It is further understood that the use of relational terms, if any, such as first and second, top and bottom, and the like are used solely to distinguish one from another entity or action without necessarily requiring or implying any actual such relationship or order between such entities or actions. Much of the inventive functionality and many of the inventive principles are best implemented with or in one or more conventional digital signal processors (DSPs), or with integrated circuits (ICs) such as custom or application specific ICs. It is expected that one of ordinary skill, notwithstanding possibly significant effort and many design choices motivated by, for example, available time, current technology, and economic considerations, when guided by the concepts and principles disclosed herein will be readily capable of programming such DSPs, or generating such ICs with minimal experimentation. Therefore, in the interest of brevity and minimization of any risk of obscuring the principles and concepts according to the present invention, further discussion of such DSPs and ICs, if any, will be limited to the essentials with respect to the principles and concepts employed by the preferred embodiments.  
     [0017] Referring initially to FIG. 8, a noise canceling apparatus  100 ,  300 ,  400  includes a Digital Phase Locked Loop (PLL)  808  that receives from an analog-to-digital converter  804 , digital I and Q representations of the a wide spectrum signal including a high amplitude noise signal. Such interfering signals may, for example be of the type commonly referred to by those skilled in the art as a “jammer signal”. Digital PLL  808  outputs a characteristic frequency, which may be, for example, the center or strongest frequency, of the jammer signal. A filter  810  is connected to receive the output of the PLL  808 , and outputs a smoothed representation of the characteristic frequency from the PLL  808 . The frequency is converted to a phase in a frequency to phase converter  812 .  
     [0018] A frequency shifter  814  shifts the I and Q signals output by the A/D converter by an amount proportional to the characteristic frequency of the interfering signal, such that the characteristic frequency is at a predetermined value. For example, the characteristic frequency may advantageously be the center frequency, and the predetermined value may be zero Hz (DC). A DC notch filter  816  removes the DC component output by the frequency shifter  814 , to remove the interfering signal. A shifter  818  receives the filtered output, and shifts this output signal, with the interfering signal removed, back to its frequency prior to shifting in shifter  814 .  
     [0019] Referring to FIG. 1, an electrical block diagram of an exemplary first embodiment of an apparatus  100  comprises an adaptive Digital Phase Locked Loop (PLL)  106 , two complex mixers  141  and  136 , a comb, i.e., sinc(f) based, filter  120 , and a DC notch filter  132 . The first embodiment  100  is an equivalent model with the same resulting functional operation as the preferred embodiment and is included for less complex conceptualization. The input to the first embodiment  100  at input nodes  102 ,  104  is a generic wide-band frequency signal plus additive narrow-band noise at a frequency of  ω 0. Note that the input signal can be viewed as a complex time varying input signal of the form I(t)+jQ(t). Alternatively, at any specific time instance, ti, the signal can be viewed as a complex valued number of the form I+jQ (i.e., I(t=ti)+jQ(t=ti)).  
     [0020] The PLL  106  comprises a conventional gain block  112  coupled to the real output  152  of a first conventional complex mixer  114 , which mixes the complex input signal with the complex output of a first conventional complex exponent function generator  108  for vector rotation. The output  118  of the gain block  112  is coupled to a conventional integrator  110 , which is coupled through a conventional negating stage  116  to generate a control signal  150  representing a phase input for the first complex exponent function generator  108 . The output  118  of the digital PLL  106  locks on to the center frequency of the narrow-band interferer.  
     [0021]FIG. 2 is a frequency-domain diagram  200  depicting the magnitude over frequency of an exemplary composite input signal  202 . As illustrated, the undesired narrow-band interfering signal is a complex noise tone  204  at a frequency of fc+25 KHz, where fc represents the carrier frequency. In other words, the interfering signal is 25 KHz above the carrier frequency. In the examples that follow, we choose a signal bandwidth of 100 KHz, a sampling rate of 1 MHz, and a carrier frequency of 200 KHz.  
     [0022] The first embodiment  100  works as follows. In spread spectrum signaling, the information-bearing signal is spread in frequency. The modulated signal therefore has a much wider bandwidth than the information-bearing signal. In addition, the spread spectrum data is treated as background noise by the digital PLL. Thus, the digital PLL  106  locks adaptively onto the center frequency of the complex tone  204 , which in this example is at carrier frequency +25 KHz. Stability of the loop is guaranteed for loop constants satisfying the equation, 0≦Ka≦2/A, where A is amplitude of the complex tone. Let T s  represent a time interval between data samples (i.e., the inverse of the sampling frequency) and {circumflex over (ω)} inst  represent the instantaneous frequency value generated by the digital PLL. The value of the signal generated at the output of the loop gain block  112  is equal to {circumflex over (ω)} inst   T   s . This signal is noisy due to the background noise associated with the spreading function of the information-bearing signal. Its DC component represents the best estimate of the center frequency of the interfering signal. For this reason, we filter this signal, preferably with a conventional M-order comb low pass Finite Impulse Response (FIR) filter  120 . Though we are not restricted in the type of low-pass filter chosen, we have chosen a comb filter due to its simplicity of implementation and its circumvention of expensive high rate multiplication operations. We note that the filter is not required to be a FIR filter, since linear phase is unnecessary. Therefore, an acceptable alternative can be a low-pass Infinite Impulse Response (IIR) filter with coefficients preferably chosen according to optimum canonical sign digit representation. The smoothed output at the filter output  122  is indicated as the filtered center frequency estimate of the interfering signal, {circumflex over (ω)} 0   T   s , in FIG. 1.  
     [0023] The estimated interference frequency is then converted to phase via a conventional first-order integrator section  124 . The integrator  124 , may for example be implemented using a twos complement binary, rollover adder such that phase limiting is also provided. The determined phase is negated in a conventional negating stage  126  so that both the negative and positive phases are output from the integrator section as control signals  128  and  130 , respectively. The negative phase represents the input argument to a second conventional complex exponent function generator  142  for vector rotation. The output of this function generator represents a second complex injection frequency, which is mixed with the original complex input signal in a second complex mixer  141 . This effectively mixes the data to baseband, centered on the frequency of the narrow-band interferer, through a lossless, complex frequency shift, thereby producing a shifted composite signal  146  in which the center frequency of the interfering signal is zero, or DC. This shifted composite signal is filtered with a conventional fixed finite impulse response (FIR) DC notch filter  132  to remove the interfering signal, thereby producing an interference-free shifted signal  148 .  
     [0024] Utilizing a fixed FIR filter response provides advantages of simplification. When the FIR direct current (DC) notch filter  132  is arranged to have a larger bandwidth than the largest interferer, the coefficients for the notch filter can be pre-computed and stored in memory. In this instance, distributed arithmetic, which circumvents the need for a multiplication operation, preferably is used to perform the notch filter operation. For the case where the notch filter bandwidth varies with signaling conditions, pre-computing several FIR filters of varying bandwidths is suggested. Computation of each filter can be likewise implemented via distributed arithmetic.  
     [0025] Finally, the interference-free shifted signal  148  is mixed in a third complex mixer  136  with a third complex injection frequency signal from a third complex exponent function generator  134  whose phase magnitude input is identical to that of the second complex exponent function generator  142 , but of opposite sign. The output nodes  138 ,  140 , coupled from the third complex mixer  136 , represent a signal in which the desired spread spectrum signal has been recovered or reconstructed with the undesired narrow-band interferer substantially removed.  
     [0026] Referring to FIG. 3, an electrical block diagram of an exemplary second embodiment of an apparatus  300  is similar to the first embodiment  100 , the essential difference being that the complex exponent function generators  108 ,  142 ,  134  and the complex mixers  114 ,  141 ,  136  have been replaced by first, second and third vector rotation elements  302 ,  304 ,  306 , which employ the well-known Coordinate Rotation Digital Computer (CORDIC) algorithms to perform the complex mixing operation. These are a class of shift-add algorithms for rotating vectors in a plane. Briefly, the CORDIC rotator performs a rotation using a series of specific incremental rotation angles selected so that each is performed by a shift and add operation. Rotation of unit vectors provides a way to compute trigonometric functions, as well as the magnitude and phase angle of an input vector. Vector rotation is also useful in many DSP applications including modulation and Fourier Transforms. A more comprehensive discussion of CORDIC techniques may be found in  A Survey of CORDIC Algorithms for FPGAs , by Ray Andraka, FPGA &#39;98,  Proceedings of the  1998  ACM/SIGDA Sixth International Symposium on Field Programmable Gate Arrays , Feb. 22-24, 1998, Monterey, Calif. pp 191-200.  
     [0027] The real output  310  of the first vector rotation element  302  is coupled to the gain block  112 . The PLL  308  also has received a new reference number to reflect the new CORDIC vector rotation element  302  therein. CORDIC advantageously is a multiplier-free algorithm as indicated above. The input to the CORDIC element is a phase, θ, and complex data of the form I+jQ. The output from the CORDIC element is a complex signal representing the vector rotation of the input signal (i.e. I+jQ&lt;θ). Therefore, by further constraining the digital PLL loop gain to be a power of two (i.e. 2 i ), no multiplication operations are required throughout the entire second embodiment  300 .  
     [0028] Those skilled in the art will recognize that the phase integration modulo operation necessary for stable implementation can be obtained by normalizing the CORDIC with an exact same rotation functionality under the phase mapping [−2π→2π] [−1 →1]. Thus, we scale the phase value by 1/2π. Note that vector rotation occurs identically to that of the second embodiment  300 , except for the conversion of radians to normalized radians. To avoid multiplication, we implement the 1/2π scaling operation by canonical sign digit shifts and adds. Note that the comb filter and digital PLL loop are now arranged to operate with the scaled frequency value of  {circumflex over (ω)}     inst     T s/2π. However in every case, the 1/2π scaling is consumed by the normalized CORDIC algorithm and does not affect the PLL loop gain or the injection frequency result. The modulo addition operation is preferably implemented by performing the additions via twos complement fraction arithmetic and by allowing the twos complement adder to overflow above 1 and below −1.  
     [0029]FIG. 4 depicts another embodiment of an interference attenuator apparatus  400 . The embodiment  400  is similar to the second embodiment  300 , the essential difference being that the first, second, and third vector rotation elements  302 ,  304 ,  306  have been replaced by first, second, and third normalized vector rotation elements  402 ,  404 ,  406 , using the [−2π→2π] [−1 →1] phase mapping described above. In addition, 1/2π phase normalization element  410  has been inserted in the PLL  408  between the gain block  112  and the output  118 .  
     [0030] Referring to FIG. 5, a frequency-domain diagram  500  depicts the magnitude over frequency of the shifted composite signal  146 . The interfering signal  204  has been moved to a predetermined frequency location, namely DC.  
     [0031] Referring to FIG. 6, a frequency-domain diagram  600  depicts the magnitude over frequency of the interference-free shifted signal  148 . The interfering signal  204  has been substantially attenuated. Referring to FIG. 7, a frequency-domain diagram  700  depicts the magnitude over frequency of recovered spread spectrum signal  702  at the output nodes  138 ,  140 . The interfering signal  204  has been effectively removed.  
     [0032] A communication device  800  is illustrated in FIG. 8. The communication device includes an antenna  801  through which communication signals are transmitted and/or received with another communication device (not shown), such as a base station, two-way communciation device, local area network, or the like. The communication device  800  includes conventional receiver RF front-end circuitry  802  for receiving and down-converting a wireless signal carrying the composite signal  202  and producing an analog output signal. The analog signal is converted to a digital signal in analog-to-digital (A/D) converter  804 . The output of the A/D converter  804  is processed in an interference removing apparatus  100 ,  300 ,  400  for removing narrower frequency, high amplitude, interfering noise signals. The interference removing apparatus may employ any of the apparatus,  100 ,  300  or  400  as described hereinabove. The output of the interference removing apparatus can be processed in conventional receiver backend circuitry  820 . Those skilled in the art will recognize that such circuitry typically includes equalizers, decoders, filters, and other elements. In typical devices, the backend circuitry must either employ very complex systems for detecting and removing the jammer signal, or the jammer signal must be tolerated. The present invention, improves over the prior art, by removing the jammer signal without requiring the complex operations.  
     [0033] The signals output from the receiver backend are input to conventional control and audio circuitry  824  of the device. It will be appreciated that the communication receiver  800  can comprise conventional elements (some of which are not shown), such as a speaker  826 , a microphone  828 , a display and/or keypad  831 , control buttons, and/or a battery, for example. A conventional transmitter  830  is provided to support two-way communications, in conjunction with the other elements of the communication device  800 .  
     [0034] Thus, it should be clear from the preceding disclosure that the present invention provides a method and apparatus for digitally processing a composite signal including a desired spread spectrum signal and an undesired narrow-band interfering signal. The method and apparatus advantageously nulls out narrow-band interfering noise within a wide-band frequency signal without the use of explicit multiplication operations or specific multiplication hardware support. According to one embodiment of the invention, only addition operations are required. Furthermore, the system adaptively adjusts to time varying frequency signals and will adjust its nulling frequency depending on the value of the undesired frequency.  
     [0035] It will be recognized by those skilled in the art that the digital PLL  808  will lock onto a single, large amplitude interference signal, so long as the narrower band interference signal has a larger amplitude than the spread spectrum signal. The interference attenuator  100 ,  300 ,  400  attenuates this one interference signal. To attenuate more than one signal, interference attenuators  100 ,  300 ,  400  can be concatenated, such that a first interference attenuator attenuates the primary interferer, and a second interference attenuator  100 ,  300 ,  400  attenuates the output of the first attenuator. To attenuate even more signals, additional interference attenuators could be used.  
     [0036] This disclosure is intended to explain how to fashion and use various embodiments in accordance with the invention rather than to limit the true, intended, and fair scope and spirit thereof. The foregoing description is not intended to be exhaustive or to limit the invention to the precise form disclosed. Modifications or variations are possible in light of the above teachings. The embodiment(s) was chosen and described to provide the best illustration of the principles of the invention and its practical application, and to enable one of ordinary skill in the art to utilize the invention in various embodiments and with various modifications as are suited to the particular use contemplated. All such modifications and variations are within the scope of the invention as determined by the appended claims, as may be amended during the pendency of this application for patent, and all equivalents thereof, when interpreted in accordance with the breadth to which they are fairly, legally, and equitably entitled.