Patent Publication Number: US-9425722-B2

Title: Electric motor control device and electric motor control method

Description:
TECHNICAL FIELD 
     The present invention relates to a control of a system where a DC voltage supplied from a battery is converted into an AC voltage using an inverter and the AC voltage is applied to an AC motor. 
     BACKGROUND ART 
     As a method of controlling a torque of an alternating current (AC) electric motor, there are known an electric current control method and a voltage control method. As the electric current control method, for example, there is known a pulse width modulation (PWM) control in which an electric current is controlled based on a vector control. As the voltage control method, there is known a rectangular wave control in which rotation of the AC electric motor is driven by applying a rectangular wave voltage. In addition, as a control for improving an output power in a so-called flux weakening area, there is known a voltage phase control in which a torque of the AC electric motor is controlled by controlling a voltage phase depending on a difference between a torque command value and an actual torque in a rectangular wave voltage control. Furthermore, there is also known a configuration capable of switching a control method depending on a situation by providing such control methods as switchable control modes. 
     However, if a steady-state torque is different between the control modes, a so-called torque discontinuity is generated when the control mode switches, and this may make a driver feel uncomfortable. In JP 2007-159368A, there is disclosed a control for suppressing the torque discontinuity at the time of control mode switching. Specifically, in the voltage phase control, a torque or an electric power is estimated, and the estimated value is fed back to the voltage phase command value. Similarly, in the current control, a torque or the like is estimated, and the estimated value is fed back to the torque command value. As a result, a steady-state torque is maintained constantly regardless of the control mode. 
     SUMMARY OF INVENTION 
     However, in the technique disclosed in JP 2007-159368A, a torque feedback loop is added to the current control mode, so that a computation load increases. Although the increase of the computation load may be alleviated by setting the computation period to be longer, control performance inevitably decreases. In addition, a higher performance computation device may be employed. However, this may increase cost. 
     In view of the aforementioned problems, the present invention has an object to provide a technology for reducing a computation load to suppress a torque discontinuity at the time of control mode switching. 
     According to an aspect of this disclosure, there is provided an electric motor control device capable of executing a current control mode in which a voltage applied to the electric motor is controlled to match a current command value generated based on a torque command value and a motor constant corresponding to a temperature and a voltage phase control mode in which a feedback operation is performed for a voltage phase based on a difference between the torque command value and a torque estimation value obtained using a torque computation formula, wherein the voltage phase control mode is selected in a high rotation area where a flux weakening control is performed. In the torque computation formula used to estimate the torque in the voltage phase control mode, the same motor constant as that used to generate the current command value in the current control mode is used. 
     Embodiments of the present invention and merits of the present invention will be described below in detail together with the attached drawings. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a control block diagram illustrating an electric motor control according to a first embodiment; 
         FIG. 2  is a control block diagram illustrating an electric motor control according to a second embodiment; 
         FIG. 3  is a control block diagram illustrating an electric motor control according to a third embodiment; 
         FIG. 4  is a control block diagram illustrating an electric motor control according to a fourth embodiment; and 
         FIG. 5  is a flowchart illustrating a control routine common to the first to fourth embodiments. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Hereinafter, an embodiment of this disclosure will be described with reference to the accompanying drawings. 
     First Embodiment 
       FIG. 1  is a control block diagram illustrating an electric motor control according to a first embodiment. This control is executed based on a flowchart described below. 
     The control according to the first embodiment is performed by switching between a current control mode and a voltage phase control mode depending on a driving state of an electric motor. The current control mode is a control mode performed to match a current command value based on a torque command value and a motor constant corresponding to a magnet temperature of an electric motor  9 . The voltage phase control mode is a control mode in which a torque feedback operation is performed for a voltage phase based on a difference between a torque command value and a torque estimation value obtained from a torque computation formula. In addition, the mode switches to the voltage phase control mode in a high rotation area where a flux weakening control is performed. 
     A current control unit  100  in  FIG. 1  includes a current command generator  1 , an interference voltage generator  2 , and a current vector controller  3 . The current control unit  100  generates d-q axis voltage command values v di*  and v qi*  for the current control mode as described below. A voltage phase control unit  200  in  FIG. 1  includes a torque calculator  14 , a torque controller  15 , and a d-q axis voltage generator  16 . The voltage phase control unit  200  generates d-q axis voltage command values v dv*  and v qv*  for the voltage phase control mode as described below. 
     The motor constant generator  18  stores a magnetic characteristic table of the electric motor  9  created in advance and outputs a magnetic flux value φ a   _   tmp  used in both the current control mode and the voltage phase control mode and a d-q axis inductance difference (Ld−Lq) tmp  as a difference between the d-axis inductance and the q-axis inductance based on a predetermined magnet temperature tmp and d-q axis current detection values i d  and i q . 
     It is noted that the predetermined magnet temperature tmp is set to a detection temperature if it is possible to detect the magnet temperature. For example, the predetermined magnet temperature tmp may be fixed to 25° C. as a performance guarantee temperature. 
     A description will now be made for the current control unit  100 . 
     Similar to the current control mode known in the art, the current command generator  1  receives a torque command value T*, a magnetic flux value φ a   _   tmp , and a d-q axis inductance difference (Ld−Lq) tmp  and outputs a d-q axis current command values i d*  and i q* . 
     The interference voltage generator  2  generates d-q axis interference voltages v d*   _   dcpl  and v q*   _   dcpl  based on the d-q axis current command values i d*  and i q* . 
     The current vector controller  3  receives the d-q axis current command values i d*  and i q*  and the d-q axis interference voltages v d*   _   dcpl  and v q*   _   dcpl , performs a vector current control utilizing a non-interference control and a current feedback control known in the art, and outputs the d-q axis voltage command values v di*  and v qi*  for the current control mode. 
     A description will now be made for the voltage phase control unit  200 . 
     The torque calculator  14  receives the magnetic flux value φ a   _   tmp , the d-q axis inductance difference (Ld−Lq) tmp , and the d-q axis current detection values i d  and i q  and calculates a torque estimation value T ca1  of the electric motor  9  based on formula (1). It is noted that a factor “p” in Formula 1 denotes the number of pole pairs of the electric motor  9 .
 
 T   cal   =p{Φ   a   _   tmp +( L   d   −L   q ) tmp   ·i   d   }i   q   (1)
 
     The torque controller  15  receives a difference between the torque command value T* and the estimated torque T cal  and outputs a value obtained by performing PI-amplification based on formula (2) as a voltage phase command value α*. It is noted that, in formula (2), “Kp” denotes a proportional gain and “Ki” denotes an integral gain. 
     
       
         
           
             
               
                 
                   
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     The d-q axis voltage generator  16  receives the voltage phase command value α*, and calculates and outputs the d-q axis voltage command values v dv*  and v qv* for the voltage phase control mode based on Formula (3). In formula (3), “V dc ” denotes a battery voltage detection value detected by the DC voltage sensor  19 , and “M*” denotes a modulation factor command value. 
     
       
         
           
             
               
                 
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     As described above, the current control unit  100  generates the d-q axis voltage command values v di*  and v qi*  for the current control mode, and the voltage phase control unit  200  generates the d-q axis voltage command values v dv*  and v qv*  for the voltage phase control mode. The control mode switching unit  17  selects any one of the command values depending on the selected control mode and outputs it. 
     Hereinafter, a description will be made for a control common to both the control modes. 
     The d-q axis/UVW-phase transformer  4  transforms the d-q axis voltage command values v d*  and v q*  into 3-phase AC voltage command values v u* , v v* , and v w*  using formula (4) based on an electric angle θ of a rotor of the electric motor  9  detected by a position detector  10 . 
     
       
         
           
             
               
                 
                   
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     The PWM converter  5  performs processes known in the art, such as a dead time compensation process or a voltage use efficiency improvement process, and generates power element driving signals D uu* , D ul* , D vu* , D vl* , D wu* , and D wl*  of the inverter  6  corresponding to the 3-phase AC command values v u* , v v* , and v w* . 
     A battery  7  is connected to the inverter  6 . The battery voltage V dc  is detected by the DC voltage sensor  19 . The inverter  6  converts the battery voltage into the pseudo sinusoidal voltages v u , v v , and v w  based on the power element driving signals D uu* , D ul* , D vu* , D vl* , D wu* , and D wl*  and outputs the result. 
     The pseudo sinusoidal voltages v u , v v , and v w  are applied to the electric motor  9 . The current detector  8  detects a U-phase current i u  and a V-phase current i v  out of overall phase currents flowing through the electric motor  9 . The W-phase current i w  that is not detected is obtained based on formula (5).
 
 i   w   =−i   u   −i   v   (5)
 
     The UVW-phase/d-q axis transformer  12  transforms the UVW-phase currents i u , i v , and i W  into the d-q axis current detection values i d  and i q  using formula (6) based on the electric angle θ of the rotor of the electric motor  9  detected by the position detector  10 . 
     
       
         
           
             
               
                 
                   
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     The rotation number calculator  11  calculates and outputs a rotation speed N of the electric motor  9  from a change rate of the electric angle θ per hour. 
     A description will now be made for the effects and advantages of the configuration described above. 
     Since the torque of the electric motor  9  is a vector product between the magnetic flux vector and the electric current vector, an absolute value of the torque of the electric motor  9  changes as a magnet temperature changes. The current is controlled constantly in the current control mode, whereas the current is not controlled in the voltage phase control mode. For this reason, a sensitivity of the torque for a change of the magnet temperature is different depending on the control mode. 
     However, even in the voltage phase control mode, similar to the method used to generate the current command value in the current control mode, the torque is estimated using a motor constant (such as the magnetic flux value φ a   _   tmp  and the d-q axis inductance difference (Ld−Lq) tmp ), and a feedback control is performed. Therefore, it is possible to improve the sensitivity of the torque as high as that of the current control mode. As a result, it is possible to reduce a difference of the steady-state torque between the current control mode and the voltage phase control mode and prevent a driver from feeling uncomfortable due to a torque discontinuity during the change of the control mode. In addition, since the torque feedback control is not executed at the time of the current control mode, it is possible to reduce a computation load. 
     Since the motor constant is calculated using the current detection value of the electric motor  9 , it is possible to generate the motor constant corresponding to the current-dependent characteristic of the inductance of the electric motor and thus improve the torque control accuracy. Furthermore, since the motor constant is computed using the magnet temperature, it is possible to suppress a change of the torque caused by a change of the temperature. 
     Second Embodiment 
       FIG. 2  is a control block diagram illustrating an electric motor control according to a second embodiment. This control is executed based on the flowchart described below. 
     According to the second embodiment, the magnet temperature tmp is set to a fixed value (25° C.), and an inductance generator  13  is provided instead of the motor constant generator  18 . As a result, the computation performed in the current control unit  100  and the voltage phase control unit  200  is different from that of the first embodiment. Hereinafter, a description will focus on a difference from the first embodiment. It is noted that the temperature “25° C.” is set based on a performance guarantee temperature or an ambient temperature of an operational environment in order to fix the magnet temperature tmp, and the invention is not limited thereto. 
     A description will now be made for the current control unit  100 . 
     The current command generator  1  receives the torque command value T*, the rotation speed N, and the battery voltage V dc  and generates and outputs the d-q axis current command values i d*  and i q*  in the case of “tmp=25° C.” by referring to a table created in advance through experiment or computation. Similarly, the interference voltage generator  2  receives the torque command value T*, the rotation speed N, and the battery voltage V dc  and generates and outputs the d-q axis interference voltage values v d*   _   dcpl , v q*   _   dcpl  in the case of “tmp=25° C.” by referring to a table created in advance through experiment or computation. It is possible to reduce a computation load by using the table in this manner. 
     A description will now be made for the voltage phase control unit  200 . 
     The torque calculator  14  receives the magnetic flux φ a   _   25° C.  at “tmp=25° C.” and the d-q axis inductance difference (Ld−Lq) 25° C.  at tmp=25° C. stored in advance and calculates an estimated torque based on these constants and the d-q axis current using formula (7).
 
 T   cal   =p{Φ   a   _   25° C. +( L   d   −L   q ) 25° C.   ·i   d   }i   q   (7)
 
     The inductance generator  13  receives the torque command value T*, the rotation speed N, and the battery voltage V dc  and generates and outputs the d-q axis inductance difference (Ld−Lq) 25° C.  by referring to a table created in advance through experiment or computation. 
     A description will now be made for the effects and advantages of the configuration described above. 
     Both the current control mode and the voltage phase control mode are based on a particular temperature condition (tmp=25° C.). Therefore, if the magnet temperature is deviated from this condition, an actual torque output is also deviated from the command value. However, a sensitivity of a torque change against a temperature change is equal between both the control modes. Therefore, it is possible to reduce a torque discontinuity at the time of the control mode switching. 
     The inductance generator  13  generates the motor constant using the torque command value T* of the electric motor  9 . Since the torque of the electric motor  9  depends on the electric current, it is possible to generate the motor constant corresponding to dependence on the electric current of the electric motor inductance and thus improve the torque control accuracy. 
     Third Embodiment 
       FIG. 3  is a control block diagram illustrating an electric motor control according to a third embodiment. This control is executed based on the flowchart described below. 
     Hereinafter, a description will focus on a difference from the second embodiment. 
     The current command generator  1  and the interference voltage generator  2  receive the torque command value T*, the rotation speed N, and the battery voltage V dc  and generates and outputs the d-q axis current command values i d*  and i q*  and the d-q axis interference voltage values v d   _   dcpl , v q*   _   dcpl  at tmp=25° C. by referring to a table created in advance through experiment or computation. It is noted that, as a pre-requisite of the table creation, a modulation rate M* in a saturation area of the motor application voltage, that is, in a flux weakening area is equally set to the modulation rate M* used when the d-q axis voltage generator  16  calculates the d-q axis voltage command values v dv*  and v qv*  for the voltage phase control mode based on the voltage phase command value α*. 
     The inductance generator  13  calculates and outputs the d-q axis inductance difference (Ld−Lq) 25° C.  using formula (8) based on a relationship between the torque command value T* as an input of the current command generator  1  and the current command values i d*  and i q*  as an output of the current command generator  1 . 
     
       
         
           
             
               
                 
                   
                     
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     That is, the inductance difference (Ld−Lq) 25° C.  is calculated using a table stored in the current command generator  1 . The table stored in the current command generator  1  essentially represents a relationship between a torque, an electric current, and a magnetic flux and thus can be used to generate the inductance difference (Ld−Lq) 25° C.    
     According to the third embodiment, the table is shared between the current command generator  1  and the inductance generator  13  as described above. Therefore, it is possible to reduce a memory capacity used to store the table. 
     Fourth Embodiment 
       FIG. 4  is a control block diagram illustrating an electric motor control according to a fourth embodiment of this disclosure. This control is executed based on the flowchart described below. 
     The fourth embodiment is similar to the second embodiment except for the method of generating the inductance difference (Ld−Lq) 25° C.  in the inductance generator  13 . 
     In the second embodiment, the inductance generator  13  generates the inductance difference (Ld−Lq) 25° C.  by referring to the table created by using the torque command value T*, the rotation speed N of the electric motor  9 , and the battery voltage V dc  as indices. However, since the inductance essentially depends on an electric current, the inductance difference (Ld−Lq) 25° C.  is generated by referring to a table created using the d-q axis current as an index according to the fourth embodiment. As a result, the torque calculator  14  estimates a torque using the inductance difference (Ld−Lq) 25° C.  obtained by referring to the magnetic flux φ a   _   25° C.  at “tmp=25° C.” stored in advance and the table representing a relationship between the electric current and the inductance stored in advance. 
     As a result, the number of indices to be referred is reduced as described above. Therefore, it is possible to simplify the computation. 
       FIG. 5  is a flowchart illustrating a control routine common to the first to fourth embodiments described above. 
     In step S 10 , the torque command value T*, the d-q axis current detection values i d  and i q , the electric angle θ, and the battery voltage detection value V dc  are obtained. In step S 20 , the control mode switching unit  17  selects any one of the voltage phase control mode and the current control mode. Here, the voltage phase control mode is selected in a high rotation area where a flux weakening control is performed, and the current control mode is selected in other areas. 
     In the case of the voltage phase control mode, the process advances to step S 30 . In step S 30 , the torque calculator  14  calculates the torque estimation value T cal  based on a predetermined magnet temperature tmp. In step S 40 , the torque controller  15  performs PI-amplification for the difference between the torque command value T* and the torque estimation value T cal  and computes the voltage phase command value α*. In step S 50 , the d-q axis voltage generator  16  computes the d-q axis voltage command values v dv*  and v qv*  from the voltage phase command value α*. 
     Meanwhile, in the case of the current control mode, the process advances to step S 60 . In step S 60 , the current command generator  1  generates the current command values i d*  and i q*  based on a predetermined temperature tmp. In step S 70 , the current vector controller  3  calculates the d-q axis voltage command values v di*  and v qi* . 
     In step S 80 , the control mode switching unit  17  sets the d-q axis voltage command values of the control mode selected in step S 20  as the voltage command values v d  and v q . In addition, the d-q axis/UVW-phase transformer  4  transforms the voltage command values v d  and v q  into the 3-phase AC voltage commands v u* , v v* , and v w* . 
     In step S 90 , based on the 3-phase AC voltage commands v u* , v v* , and v w* , the 3-phase AC voltage is applied to the electric motor  9  using the PWM converter  5  and the inverter  6 . 
     Through steps S 30  and S 60  described above, it is possible to suppress a torque discontinuity at the time of the control mode switching without adding a new configuration to the current control mode known in the art. 
     Although various embodiments of this disclosure have been described hereinbefore, they are just for illustrative purposes and are not intended to specifically limit the technical scope of the invention. Instead, it would be appreciated that various changes or modifications may be possible without departing from the spirit and scope of the invention. 
     This application claims priority based on Japanese Patent Application No. 2012-057741 filed with the Japan Patent Office on Mar. 14, 2012, the entire contents of which are expressly incorporated herein by reference.