Patent Publication Number: US-8120536-B2

Title: Antenna isolation

Description:
RELATED APPLICATION INFORMATION 
     The present application claims the benefit under 35 U.S.C. §119(e) of the priority date of U.S. Provisional Patent Application Ser. No. 61/044,382 filed Apr. 11, 2008, the entire contents of which are hereby expressly incorporated by reference. 
    
    
     TECHNICAL FIELD 
     The present invention relates to a dual polarized antenna element and an antenna array, in which the antenna element includes:
         a first feeder for feeding the antenna element in a first polarization direction, and   a second feeder for feeding the antenna element in a second polarization direction.       

     BACKGROUND OF THE INVENTION 
     Dual polarised or X-polarised antennas are today commonly used in cellular systems for mobile communication. The use of such antennas allows the use of polarisation diversity techniques to combat signal fading in the system. Compared to the use of vertical polarised antennas and space diversity techniques the number of antennas needed is reduced to half, which saves costs and reduces the size and the visual appearance of the antenna installations. 
     One important performance measure for dual polarised antennas is the isolation between the two antenna ports feeding the two polarisations. Typically, an isolation of more than 30 dB between the ports is wanted, which corresponds to a power coupling of less than 1/1000 between the ports. 
     An aperture coupled patch antenna element is a commonly used antenna type for dual polarised systems. In aperture coupled patch antenna elements, one or more metallic patches are fed by a micro strip feeding arrangement through a cross shaped aperture in a ground plane, as is shown in  FIG. 1 . Here, the antenna element  101  includes a radiating patch  103 , fed through an aperture  109  by a microstrip feed line  105  positioned between a shielding cage  102  and a printed circuit board. 
     Isolation between a transmitting and a receiving signal path in a dual polarized antenna has been described in, for instance, prior art document U.S. Pat. No. 6,509,883. According to this document, a signal being transmitted from a first antenna element having one polarisation is received by a second antenna element having another polarisation, thereby causing an unwanted signal to be received by the second antenna element. In order to compensate for this, a compensation path is arranged between the transmitting and receiving signal paths, where the compensation path has a length such that the compensation signal travelling through the compensation path and the unwanted signal have equal magnitude and opposite phase when they meet in the receiving signal path. 
     Prior art solutions like the one described in U.S. Pat. No. 6,509,883, have a disadvantage in that they only compensate for signals having been transmitted from one antenna element and received by another antenna element. Thus, no solution is shown for solving the problem of capacitive coupling related to the feeders themselves. 
     In U.S. Pat. No. 6,509,883, the compensation path as well as the transmitting and receiving signal paths have to be adapted to have certain lengths in order to be able to cancel out the unwanted signal, having been transmitted from the first antenna and received by the second antenna, since a difference in length of an odd number of half wavelengths has to be present between the paths traveled by the unwanted signal and the compensation signal. 
     The prior art solution will therefore only cancel out this specific unwanted signal. Other unwanted signals, resulting from couplings other than this one, such as unwanted signals originating from capacitive coupling between the feeders in a point where the feeders are close to each other, will not be cancelled by the solution shown in this document, since the distinctive length requirements of the signal paths result in cancellation of the unwanted signal only if the unwanted signal and the compensation signal have traveled exactly those required lengths. 
     Also, a capacitive coupling between the feeders may take place at a very unfortunate point, for which a difference in length of an even number of half wavelength results between the paths traveled by the unwanted signal and by the compensation signal in U.S. Pat. No. 6,509,883. The compensation signal would in this case add to the unwanted signal instead of cancelling it. 
     Further, due to the signal path length requirements, the antenna element shown in this document has to have a certain size to achieve efficient cancellation, which is disadvantageous. 
     Thus, there is a problem in prior art relating to cancellation of different kinds of couplings being present in a dual polarized antenna element. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide a dual polarised antenna element that solves the above stated problems. 
     The present invention aims to provide a dual polarised antenna element, which offers improved antenna isolation for all kinds of essentially capacitive couplings between the feeders. The present invention thus aims to provide compensation for capacitive coupling between the feeders, also including a capacitive coupling occurring via the radiating part, for example a radiating patch, of the antenna element. 
     According to an embodiment of the present invention, the object is for a dual polarized antenna element achieved by the use of:
         a compensation line being arranged between the first and the second feeders for compensating for an imbalance caused by an essentially capacitive coupling between the first and second feeders, where   the compensation line is connected to the first and second feeders in close proximity to a radiating part of the antenna element, and has a short electrical length θ and a high impedance relative to an impedance of the first and second feeders, respectively, thereby giving the compensation line an essentially inductive character.       

     The object is also achieved by an antenna array including at least two such dual polarized antenna elements. 
     Thus, the present invention achieves compensation of mutual coupling in dual polarized antenna elements using a compensation line being connected between the input ports. When this compensation line is short in relation to the wavelength, this connection will act as an inductive element well suited to compensate for the capacitive mutual coupling in the antenna element. 
     The dual polarised antenna element according to the present invention has the advantage that it can provide good antenna isolation through an efficient compensation for essentially all types of capacitive coupling between the feeders in the antenna element, including capacitive coupling between the feeders and the radiating part of the antenna element. The compensation is achieved by the use of a compensation line, which is small in size, not costly to produce, easy to implement and which efficiently cancels out the capacitive coupling being present by its inductive character. 
     According to an embodiment of the present invention, the dual polarized antenna element is of the aperture coupled patch antenna type. Each feeder here includes a pair of feed lines extending along slots of a cross shaped aperture such that the feed lines cross each other at a mutual distance, resulting in a capacitive coupling between the feeders. Such a crossing can be arranged as an air-bridge. In the antenna element according to this embodiment, this capacitive coupling is cancelled by the high impedance connection between the feeders. 
     Detailed exemplary embodiments and advantages of the antenna elements and antenna arrays of the present invention will now be described with reference to the appended drawings illustrating some preferred embodiments. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a prior art aperture coupled patch antenna element. 
         FIG. 2  shows an unbalanced prior art antenna element. 
         FIGS. 3   a - b  show unbalanced prior art antenna elements. 
         FIGS. 4   a - c  schematically show dual polarized antenna elements according to the present invention. 
         FIG. 5  schematically illustrates mutual coupling. 
         FIGS. 6   a - b  schematically illustrates capacitive mutual coupling. 
         FIGS. 7   a - b  illustrates transmission line impedances. 
         FIGS. 8   a - b  show simulations for a prior art antenna element (a), and for an antenna element according to the present invention (b). 
         FIGS. 9   a - b  show simulations for a prior art antenna element (a), and for an antenna element according to the present invention (b). 
         FIGS. 10   a - b  show simulations for a prior art antenna element (a), and for an antenna element according to the present invention (b). 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Dual polarized antenna elements commonly suffer from imbalance due to mutual coupling for various reasons. Even though an antenna element may show a geometrical symmetry to a large extent, including the radiating part and the majority of the feed network, we typically have one or more points of asymmetry causing mutual coupling. 
       FIG. 2  shows one example of this for a patch antenna element including a ground plane  202 , a top patch  203  and a lower patch  204 . Here, an electromagnetically coupled patch element is fed by two orthogonal feeders  205 ,  206 , both with a capacitive coupling to the two stacked patches. The antenna element is here not symmetrical, since the feeder connections are not symmetrical. For example, if we look into the element along for example the feeder  205  at the bottom of the figure, we see that only one side (the left side) of the other sides of each patch is loaded by another feeder  206 , while the other sides (for instance the right side) have an open circuit. Thus, the antenna element is not symmetrical around the plane of the dashed line  207 , since there is no feeder connection at the right side of the antenna element. 
     In  FIG. 3   a , and more in detail in  FIG. 3   b , an aperture coupled patch antenna element  301  having a shielding cage  302  for back radiation and a cross-shaped aperture  309  is disclosed. Here, each of the feeders  305 ,  306 , feeding a polarization, respectively, includes a pair of feed lines  307 ,  308  extending in parallel along the cross shaped aperture  309 , respectively, such that a two of those feed lines cross each other in one point  310 . Because of the symmetrical shape of the micro strip feeders, including the feed lines, each feeding one polarisation, they need to cross each other in at least one point  310 , as can be seen in  FIGS. 3   a  and  3   b . This at least one crossing  310  is typically achieved by using an air bridge for one of the polarizations. This air bridge crossing destroys the symmetry of the antenna element and imposes a capacitive coupling between the two feeders  305 ,  306 . 
     Thus, in both of the cases shown in  FIGS. 2 and 3 , there is an asymmetry present, which will cause mutual port-to-port coupling between the port P 1  and the port P 2  of the feeders. This mutual coupling and its corresponding imbalance have to be mitigated in order to achieve efficient antenna isolation. 
     According to the present invention, as will be described more in detail below, it has been discovered that such mutual coupling between the feeders often is of essentially capacitive character. From this finding, it has further been realized that an element having an essentially inductive character connected between the feeders could be used for reducing the mutual coupling between the feeders. 
     In  FIGS. 4   a - 4   c , three different types of dual polarized antenna elements according to different embodiments of the present invention are shown schematically. (Reference numbers are here only given to parts that are needed for explanation of the present invention.) These antenna elements  401  are dual polarized antenna elements and include a first feeder  405  for feeding said antenna element  401  in a first polarization direction. The first feeder  405  has a connection port P 1 . The antenna elements  401  further have a second feeder  406  for feeding said antenna element  401  in a second polarization direction, also being provided with a connection port P 2 . 
       FIG. 4   a  schematically illustrates a general dual polarized antenna element  401 , being fed by two feeders  405 ,  406 , having mutual coupling between them. 
     As shown in  FIG. 4   b , for the case that the antenna element  401  is an aperture coupled patch antenna element having a cross-shaped aperture, each one of the feeders  405 ,  406  includes a pair of feed lines  407 ,  408  extending in parallel along the cross shaped aperture  409 , on each side thereof, respectively, such that two of those feed lines  407 ,  408  cross each other in one point  410 , typically being arranged as an air bridge. Such an antenna structure could also result in more than one crossing of feed lines, depending on the shape of the feed lines. 
     According to the present invention, in order to compensate for the imbalance resulting from the mutual coupling between the feeders, a compensation line  420  is arranged between said first and said second feeders  405 ,  406 . The compensation line  420  should be connected to the first and second feeders  405 ,  406  in a point on each of the feeders that is in close proximity to a radiating part of the antenna element. 
     As was stated above (and will be proven below), the mutual coupling between the feeders is of an essentially capacitive character and can be cancelled by the compensation line  420 , if the compensation line  420  has an essentially inductive character. This is, according to the present invention, achieved by arranging the compensation line  420  such that its electrical length θ is short and that it is thin such that it has high impedance relative to an impedance of the first and second feeders  405 ,  406 . These characteristics of the compensation line  420  make the compensation line essentially inductive. 
     More in detail, as will be shown below, in order to achieve an inductive character for the compensation line  420 , the electrical length θ of the compensation line  420  should be small, preferably being less than 2π/3 rad, thus θ&lt;2π/3 rad. However, as is clear to a skilled person, also other lengths than this could be advantageous for different implementations. 
     Also, the compensation line  420  should have an impedance that is at least twice as high as the impedance for the feeders  405 ,  406 . The electrical length θ is, as is well known for a person skilled in the art, a length that is related to the wavelength of the signal being transmitted. 
     Thus, by the compensation line  420  according to the present invention, being connected between the first and second feeders  405 ,  406 , a novel method of coupling the polarisations together via an essentially inductive connection is used, in such way that the magnitude and phase of this coupling cancels the mutual coupling in other parts of the antenna element. Thereby, a required isolation level is achieved at low cost, which is small in size and easy to implement. 
     In  FIG. 4   c , for a dual polarized patch antenna, the compensation line  420  is implemented by a high impedance microstrip line in close proximity of the radiating patch  403 . In order to have an inductive character, the compensation line  420  should have a short electrical length θ and have an impedance, which is much higher than the impedance for the feeders. For example, the feeders  405 ,  406  can have an impedance of around 50Ω, whereas the compensation line has an impedance of around 220Ω. 
     The compensation line is connected to the first feeder  405  at a first distance D 1  from the radiating part of the antenna element, for instance a radiating patch. The compensation line is also connected to the second feeder  406  at a second distance D 2  from the radiating part. According to an embodiment of the present invention, the first and second distances should be very short relative to the wavelength of the transmitted signal. The first and second distances should preferably be much less than half of the wavelength of the transmitted signal, and more preferably much less than a quarter of the wavelength of the transmitted signal, in order to efficiently cancel the capacitive coupling between the feeders. Thus, preferably D 1 &lt;&lt;λ/2 and D 2 &lt;&lt;λ/2, and more preferably D 1 &lt;&lt;λ/4 and D 2 &lt;&lt;λ/4. 
     By the use of such a compensation line, having an inductive character, the capacitive coupling between the feeders is cancelled, as will be shown in the following. 
     Such a capacitive coupling can occur in any situation where a feeder or a feed line of one polarization is close to a feeder or a feed line of another polarization. Such a situation can thus occur in an air-bridge, but also somewhere else in the antenna element, where feeders run in close distance to each other. Also, as is exemplified below, there can be a capacitive coupling between one or both of the feeders and the radiating part of the antenna. 
     It will now be shown that a mutual coupling between the feeders, including coupling between the feeders and the radiating parts of the two polarizations, often is of capacitive character and that this mutual coupling can be cancelled by the use of a compensation line between the feeders having an essentially inductive character. 
     A general description of mutual coupling in a radiating part is shown in  FIG. 5 . An antenna element with two input ports is represented by a scattering matrix S or by an impedance matrix Z, both being of the dimension 2×2. Each port here corresponds to one of the two orthogonal polarizations of the radiated wave. 
     The scattering matrix S provides the relationship between ingoing voltage waves (plus sign) and outgoing voltage waves (minus sign) on the ports:
 
 V   −   =SV   +   (1)
 
     The impedance matrix Z determines the ratio between voltage vector V and current vector I on the lines:
 
 V=ZI   (2)
 
     If all ports have the same characteristic impedance Z 0 , these are related by the following well-known matrix equation:
 
 Z=Z   0 ( E+S )( E−S ) −1  
 
 S= ( Z+Z   0   E ) −1 ( Z−Z   0   E )′  (3)
 
where E is the identity matrix.
 
     In particular, from the matrix equation (3) it follows that the mutual coupling between the two ports  1  and  2 , S 21 , is related to the mutual impedance as: 
     
       
         
           
             
               
                 
                   
                     S 
                     21 
                   
                   = 
                   
                     
                       2 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         Z 
                         21 
                       
                       ⁢ 
                       
                         Z 
                         0 
                       
                     
                     
                       
                         
                           ( 
                           
                             
                               Z 
                               11 
                             
                             + 
                             
                               Z 
                               0 
                             
                           
                           ) 
                         
                         ⁢ 
                         
                           ( 
                           
                             
                               Z 
                               22 
                             
                             + 
                             
                               Z 
                               0 
                             
                           
                           ) 
                         
                       
                       - 
                       
                         
                           Z 
                           12 
                         
                         ⁢ 
                         
                           Z 
                           21 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
     Further, in  FIG. 5  we have a second 2×2 matrix defined by S M  or Z M . When analyzing  FIG. 5 , it is clear that we, in general, can design a loss-less matrix S M  such that the coupling from port  1 ′ to  2 ′ is zero. This could be done by using, e.g., a directive coupler. 
     In accordance with the present invention, we will here study a special case of cross-polar coupling in the antenna element, which is the case when this coupling is a result of a capacitance between the feeders and the radiating parts of the two polarizations. This is illustrated in  FIGS. 6   a  and  6   b.    
     In general, the mutual coupling often includes capacitive coupling between at least one of the first and second feeders and the radiating part, here being a patch, of said antenna element. 
       FIG. 6   a  shows an antenna element defined by a matrix Z with mutual coupling represented by a capacitance C. Note that the ground reference line in  FIG. 5  here has been removed for clarity.  FIG. 6   a  also shows a compensation connection in the form of an inductance L, in accordance with the present invention. 
       FIG. 6   b  shows the antenna element from  FIG. 6   a , but with the two shunt loads, corresponding to the mutual coupling and the compensation connection, being represented by a single load
   jX=jωL+ 1 /jωC,    
and Z′ being replaced by Z.
 
     Here, the elements of the impedance matrix Z can be determined from circuit theory as: 
                     Z   11     =       Z   22     =           V   1       I   1       ⁢     |       I   2     =   0         =         Z   0     //     (       Z   0     +     j   ⁢           ⁢   X       )       =         Z   0   2     +     j   ⁢           ⁢     XZ   0             2   ⁢     Z   0       +     j   ⁢           ⁢   X                       (   5   )               
and by performing voltage division and (5):
 
     
       
         
           
             
               
                 
                   
                     
                       
                         
                           Z 
                           12 
                         
                         = 
                           
                         ⁢ 
                         
                           Z 
                           21 
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             
                               V 
                               2 
                             
                             
                               I 
                               1 
                             
                           
                           ⁢ 
                           
                             | 
                             
                               
                                 I 
                                 2 
                               
                               = 
                               0 
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             
                               
                                 Z 
                                 0 
                               
                               
                                 ( 
                                 
                                   
                                     Z 
                                     0 
                                   
                                   + 
                                   
                                     j 
                                     ⁢ 
                                     
                                         
                                     
                                     ⁢ 
                                     X 
                                   
                                 
                                 ) 
                               
                             
                             ⁢ 
                             
                               
                                 V 
                                 1 
                               
                               
                                 I 
                                 1 
                               
                             
                           
                           ⁢ 
                           
                             | 
                             
                               
                                 I 
                                 2 
                               
                               = 
                               0 
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             
                               Z 
                               0 
                             
                             ⁡ 
                             
                               ( 
                               
                                 
                                   Z 
                                   0 
                                   2 
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   
                                     XZ 
                                     0 
                                   
                                 
                               
                               ) 
                             
                           
                           
                             
                               ( 
                               
                                 
                                   Z 
                                   0 
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   X 
                                 
                               
                               ) 
                             
                             ⁢ 
                             
                               ( 
                               
                                 
                                   2 
                                   ⁢ 
                                   
                                     Z 
                                     0 
                                   
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   X 
                                 
                               
                               ) 
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             
                               Z 
                               0 
                               2 
                             
                             
                               
                                 2 
                                 ⁢ 
                                 
                                   Z 
                                   0 
                                 
                               
                               + 
                               
                                 j 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 X 
                               
                             
                           
                           . 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
     Substitution of (5-6) in (4) gives: 
     
       
         
           
             
               
                 
                   
                     
                       
                         
                           S 
                           21 
                         
                         = 
                           
                         ⁢ 
                         
                           
                             
                               2 
                               ⁢ 
                               
                                 Z 
                                 0 
                                 2 
                               
                             
                             
                               ( 
                               
                                 
                                   2 
                                   ⁢ 
                                   
                                     Z 
                                     0 
                                   
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   X 
                                 
                               
                               ) 
                             
                           
                           ⁢ 
                           
                             
                               ( 
                               
                                 
                                   2 
                                   ⁢ 
                                   
                                     Z 
                                     0 
                                   
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   X 
                                 
                               
                               ) 
                             
                             
                               
                                 
                                   ( 
                                   
                                     
                                       Z 
                                       0 
                                       2 
                                     
                                     + 
                                     
                                       j 
                                       ⁢ 
                                       
                                           
                                       
                                       ⁢ 
                                       
                                         XZ 
                                         0 
                                       
                                     
                                     + 
                                     
                                       
                                         Z 
                                         0 
                                       
                                       ⁡ 
                                       
                                         ( 
                                         
                                           
                                             2 
                                             ⁢ 
                                             
                                               Z 
                                               0 
                                             
                                           
                                           + 
                                           
                                             j 
                                             ⁢ 
                                             
                                                 
                                             
                                             ⁢ 
                                             X 
                                           
                                         
                                         ) 
                                       
                                     
                                   
                                   ) 
                                 
                                 2 
                               
                               - 
                               
                                 Z 
                                 0 
                                 4 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             2 
                             ⁢ 
                             
                               
                                 Z 
                                 0 
                                 3 
                               
                               ⁡ 
                               
                                 ( 
                                 
                                   
                                     2 
                                     ⁢ 
                                     
                                       Z 
                                       0 
                                     
                                   
                                   + 
                                   
                                     j 
                                     ⁢ 
                                     
                                         
                                     
                                     ⁢ 
                                     X 
                                   
                                 
                                 ) 
                               
                             
                           
                           
                             
                               ( 
                               
                                 
                                   3 
                                   ⁢ 
                                   
                                     Z 
                                     0 
                                     2 
                                   
                                 
                                 + 
                                 
                                   j 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   2 
                                   ⁢ 
                                   
                                     XZ 
                                     0 
                                   
                                 
                               
                               ) 
                             
                             - 
                             
                               Z 
                               0 
                               4 
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                         ⁢ 
                         
                           
                             
                               
                                 Z 
                                 0 
                               
                               ⁡ 
                               
                                 ( 
                                 
                                   
                                     2 
                                     ⁢ 
                                     
                                       Z 
                                       0 
                                     
                                   
                                   + 
                                   
                                     j 
                                     ⁢ 
                                     
                                         
                                     
                                     ⁢ 
                                     X 
                                   
                                 
                                 ) 
                               
                             
                             
                               
                                 4 
                                 ⁢ 
                                 
                                   Z 
                                   0 
                                   2 
                                 
                               
                               + 
                               
                                 j 
                                 ⁢ 
                                 
                                     
                                 
                                 ⁢ 
                                 6 
                                 ⁢ 
                                 
                                   XZ 
                                   0 
                                 
                               
                               - 
                               
                                 2 
                                 ⁢ 
                                 
                                   X 
                                   2 
                                 
                               
                             
                           
                           . 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
     Equation (7) shows that, in order to have zero coupling when X is real, we need to have X→∞. 
     Since jX is a parallel circuit we have: 
     
       
         
           
             
               
                 
                   
                     j 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     X 
                   
                   = 
                   
                     
                       1 
                       
                         
                           j 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           ω 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           C 
                         
                         + 
                         
                           1 
                           
                             j 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             ω 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             L 
                           
                         
                       
                     
                     = 
                     
                       
                         j 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         ω 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         L 
                       
                       
                         1 
                         - 
                         
                           
                             ω 
                             2 
                           
                           ⁢ 
                           LC 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   8 
                   ) 
                 
               
             
           
         
       
     
     Note here that, from a feeder input port point of view, the capacitive mutual coupling and the compensation line together form a parallel resonance circuit. 
     Thus, the solution is the well-known resonance condition: 
     
       
         
           
             L 
             = 
             
               
                 
                   1 
                   
                     
                       ω 
                       2 
                     
                     ⁢ 
                     C 
                   
                 
                 ⇒ 
                 
                   X 
                   → 
                   
                     ∞ 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     and 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     
                       S 
                       21 
                     
                   
                 
               
               = 
               0. 
             
           
         
       
     
     Therefore, the mutual coupling can be cancelled by the use of a compensation line between the feeders having an inductive character. 
     In the following, it will be shown that this inductive compensation line can be implemented as a connection between the feeders having a short electrical length and being thin, such that it has a high impedance in relation to the feeder impedance. 
     We have seen above that mutual coupling from a capacitance can be compensated by adding an inductive element between the feeders. At microwave frequencies (e.g. above 1 GHz), this is preferably done by using for example a transmission line rather than discrete components. An illustration of the use of such a transmission line is shown in  FIGS. 4   a - c.    
     Since the characteristic impedance of a transmission line is 
                 Z   c     =       L   C         ,         
a high impedance transmission line should correspond to a large inductance.
 
     The question is then in which sense such a thin transmission line may be seen as the discrete element required by equation (7) above. Consider the transmission line shown in  FIG. 7 . In  FIG. 7   a , a high impedance transmission line of electrical length θ is connected to a line with the system impedance Z 0 . In  FIG. 7   b , a general case is shown. 
     The input impedance Z′ at the beginning of the high impedance line is related to the impedance of the load Z L  by the well-known transmission line formula: 
     
       
         
           
             
               
                 
                   
                     Z 
                     ′ 
                   
                   = 
                   
                     
                       Z 
                       m 
                     
                     ⁢ 
                     
                       
                         
                           Z 
                           L 
                         
                         + 
                         
                           j 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           
                             Z 
                             m 
                           
                           ⁢ 
                           tan 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           θ 
                         
                       
                       
                         
                           Z 
                           m 
                         
                         + 
                         
                           j 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           
                             Z 
                             L 
                           
                           ⁢ 
                           tan 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           θ 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   9 
                   ) 
                 
               
             
           
         
       
     
     If the high impedance transmission line is short, i.e. θ&lt;&lt;1 rad, we may approximate equation (9) as: 
                         Z   ′     ≈       Z   m     ⁢         Z   L     +     j   ⁢           ⁢     Z   m     ⁢   θ           Z   m     +     j   ⁢           ⁢     Z   L     ⁢   θ             ⁢     
     ⁢           =         Z   m     ⁢           Z   L     ⁢     Z   m       +     j   ⁢           ⁢     (       Z   m   2     +     Z   L   2       )     ⁢   θ     +       Z   L     ⁢     Z   m     ⁢     θ   2             Z   m   2     +       Z   L   2     ⁢     θ   2             ⁢     
     ⁢           ≈       Z   L     +     j   ⁢           ⁢   θ   ⁢         Z   m   2     +     Z   L   2         Z   m               ,           (   10   )               
where we have used tan θ≈ sin θ≈θ and then dropped the θ 2 -terms. From equation (10), it is clear that the effect of a short high impedance line is to add a positive series reactance. If the line is very thin so that the impedance is very high, the total impedance is simply:
 
 Z′≈Z   L   +jZ   m θ  (11)
 
     Thus, by connecting a compensation line between the feeders, an inductive element between the feeders is added, if the compensation line has a short electrical length θ and a high impedance in relation to the impedance of the feeders. 
     Thus, as was deducted above, such a high impedance inductive compensation line cancels the mutual coupling between the feeders. High impedance here means high impedance relative to the impedance of the feeders used for feeding the polarizations. 
     In connection with equation 10 above, it is, for pedagogic reasons, stated that the electrical length θ of the compensation line should be much less than 1 rad, in order to a result in an approximated expression. However, for practical implementations, according to one embodiment of the invention, the electrical length θ should preferably be less than 2π/3 rad, thus θ&lt;2π/3 rad. This electrical length also results in a compensation line having an essentially inductive character. 
     Also, as is clear for a skilled person studying equations 10-11 and  FIG. 5 , different electrical lengths θ of the compensation line, having an essentially inductive character, can be suitable for different implementations of the invention. Therefore, according to an embodiment of the present invention, the electrical length θ of the essentially inductive compensation line is longer than 2π/3 rad. 
     As non-limiting numerical examples, the feeders can have an impedance of 50Ω, and the compensation line can have an impedance of more than twice the feeder impedance, for instance 220Ω. The compensation line can, for instance, be implemented as a 0.5 mm wide microstrip line. Further, the patches can have a size of, for instance, 66 mm or 56 mm. 
     The antenna element of the present invention has been designed and simulated for signals in the frequency interval 1800 MHz to 2200 MHz. The inventive idea of the present invention may, however, also be implemented in other frequency intervals, as is clear to a skilled person. 
     Further, according to an embodiment of the present invention, dual polarised antenna elements of the present invention are arranged in an antenna array. Here, the two polarisations of two patches of two antenna array elements are each fed by a first feeder and a second feeder. According to the embodiment of the invention, there is arranged a compensation line between the first and second feeders in close proximity of each of the patches, respectively, thereby enhancing the antenna isolation of the antenna elements of the array. As is clear to a skilled person, such an antenna array can include essentially any number of dual polarized antenna elements according to the present invention. 
     Also, according to an embodiment of the present invention, the antenna isolation of the present invention is combined with other techniques for improving antenna isolation, being any one of the techniques of parasitic impedances and/or shield wall and/or asymmetrical/rectangular patches and/or diagonal apertures and/or shifted feed positions. Such a combination has the advantage of even further enhancing the level of isolation. 
     As is obvious for someone skilled in the art, the present invention can be used on essentially any dual polarised antenna element, although, for illustrational reasons, it is mainly described in terms of patch antennas, such as aperture coupled patch antennas, in this specification. 
       FIGS. 8-10  show simulations of coupling, reflection and radiation patterns for a dual polarised patch antenna element according to prior art and according to the present invention.  FIGS. 8   a ,  9   a  and  10   a  show simulations for a prior art antenna, basically an antenna element as the one shown in  FIG. 2 .  FIGS. 8   b ,  9   b , and  10   b  show simulations for an antenna element according to the present invention, more specifically for an antenna element as the one shown in  FIG. 4   c , having a compensation line arranged between the feeders. 
     In these simulations, a microstrip line has been used as the compensation line  420 , the microstrip line being implemented as a 0.5 mm wide line resulting in an impedance of 220Ω for the compensation line  420 . The first and second feeders  205 ,  206 ,  405 ,  406  feeders here have an impedance of 50Ω. Thus, a current division between the 50Ω impedance of the first and second feeders  405 ,  406  and the 220Ω impedance of the compensation line  420  will take place in the antenna element according to the present invention. 
     As can be seen in  FIGS. 8   a  and  8   b , the mutual coupling  830  is much lower for the antenna element of the present invention (shown in  FIG. 8   b ), as for the prior art antenna element (shown in  FIG. 8   a ). Note here that the two diagrams have differing scales. The antenna element of the present invention thus has a coupling being around 30 dB between the feeder ports. Also, the reflection  840  is more or less similar for the prior art antenna element and the antenna element of the present invention. 
     Further,  FIGS. 9   a  and  9   b  show a simulated radiation pattern at 2000 MHz for the azimuth plane (φ=0° in the coordinate system shown in  FIG. 4   c ) for the prior art antenna element ( FIG. 9   a ) and for the antenna element of the present invention ( FIG. 9   b ), both being simulated as having infinite ground planes. 
     As can be seen in  FIGS. 9   a  and  9   b , the cross polarisation, E_cross, is greatly improved for the antenna element according to the present invention ( FIG. 9   b ), as compared to the prior art antenna element ( FIG. 9   a ). For the present invention, the level of the cross polarisation is 30 dB on the z-axis (THETA=0), which is very desirable. THETA is here defined as the angle from a z-axis being perpendicular to both the x-axis and y axis in the system of coordinates defined in  FIG. 4   c.    
     The radiation pattern in the direction of the polarisation, E_co, is very similar for both the prior art antenna element ( FIG. 9   a ) and for the antenna element of the present invention ( FIG. 9   b ). This tells us that that we have not deteriorated that characteristic of the radiation at the same time as we have gained a lot for the cross polarisation. 
       FIGS. 10   a  and  10   b  show a simulated radiation pattern at 2000 MHz for the E-plane (φ=45° in the coordinate system shown in  FIG. 4   c ) for the prior art antenna element ( FIG. 10   a ), and for the antenna element of the present invention ( FIG. 10   b ), both being simulated as having infinite ground planes. 
     As for the azimuth plane, it can be seen in  FIGS. 10   a  and  10   b , that the cross polarisation, E_cross, is greatly improved for the antenna element according to the present invention, as compared to the prior art antenna element. A very good isolation level of 30 dB on the z-axis (THETA=0) for the cross polarisation is here also achieved for the present invention. 
     The radiation pattern in the direction of the polarisation, E_co, is also here not deteriorated by the compensation line of the present invention. 
     Further, in corresponding simulations for an antenna array, including two antenna elements according to the present invention, the coupling isolation (E_cross) for the radiation pattern for the antenna array has shown to be more than 23 dB.