Patent Publication Number: US-2023155533-A1

Title: Motor control device, electric vehicle, and motor control method

Description:
TECHNICAL FIELD 
     The present invention relates to a motor control device, an electric vehicle, and a motor control method. 
     BACKGROUND ART 
     A control device that includes inverters of two systems corresponding to two sets of multiphase winding sets and controls energization to each winding set for a multiphase AC motor including the two sets of multiphase winding sets in a control device of a motor is known. 
     PTL 1 discloses a device as follows. The device includes inverters of two system and a control unit. The inverters of the two systems are provided electrically independently to correspond to two sets of multiphase winding sets that form a stator of a multiphase AC motor and apply a rotating magnetic field to a rotor, and output an AC voltage to the two sets of multiphase winding sets. The control unit controls a phase difference of the AC voltage applied to the two sets of multiphase winding sets. The control unit sets a control range including a reference phase difference capable of reducing a harmonic component of a specific order with respect to the phase difference and changes the phase difference in the control range based on required characteristics of the multiphase AC motor or so as to cause fluctuation in energization of the multiphase AC motor. 
     CITATION LIST 
     Patent Literature 
     PTL 1: JP 2015-213407 A 
     SUMMARY OF INVENTION 
     Technical Problem 
     The above-described device disclosed in PTL 1 cannot sufficiently suppress a vibration and noise generated in the motor. 
     Solution to Problem 
     According to the present invention, a motor control device includes a first inverter circuit and a second inverter circuit of a redundant system, the first inverter circuit and the second inverter circuit controlling a motor, and a control unit that controls the first inverter circuit and the second inverter circuit. The first inverter circuit converts the DC power into the AC power based on a PWM signal generated by using a first carrier signal. The second inverter circuit converts the DC power into the AC power based on a PWM signal generated by using a second carrier signal. The control unit shifts phases of the first carrier signal and the second carrier signal by using, as a reference, pulsation of an electromagnetic force caused by a magnetic circuit of the motor. 
     According to the present invention, there is provided a motor control method in a motor control device including a first inverter circuit and a second inverter circuit of a redundant system, the first inverter circuit and the second inverter circuit controlling a motor, and a control unit that controls the first inverter circuit and the second inverter circuit. The motor control method includes converting, by the first inverter circuit, the DC power into the AC power based on a PWM signal generated by using a first carrier signal, converting, by the second inverter circuit, the DC power into the AC power based on a PWM signal generated by using a second carrier signal, and shifting, by the control unit, the phases of the first carrier signal and the second carrier signal by using, as a reference, pulsation of an electromagnetic force caused by a magnetic circuit of the motor. 
     Advantageous Effects of Invention 
     According to the present invention, it is possible to suppress a vibration and noise generated in a motor. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG.  1    is an overall configuration diagram of a motor drive system including a motor control device. 
         FIGS.  2 (A),  2 (B),  2 (C) , and  2 (D) are diagrams illustrating torque ripples in a case where the present embodiment is not applied. 
         FIGS.  3 (A),  3 (B),  3 (C) , and  3 (D) are diagrams illustrating torque ripples in a case where the present embodiment is applied. 
         FIGS.  4 (A) and  4 (B)  are diagrams illustrating motor pulsation maps. 
         FIGS.  5 (A) and  5 (B)  are diagrams illustrating circumferential carrier phase maps. 
         FIGS.  6 (A) and  6 (B)  are diagrams illustrating radial carrier phase maps. 
         FIG.  7    is a diagram illustrating a relationship between a rotation speed of a motor and an excitation frequency. 
         FIGS.  8 (A) and  8 (B)  are diagrams illustrating a frequency of a voltage command and a carrier frequency fc. 
         FIG.  9    is a flowchart illustrating processing of a control unit in the motor control device. 
         FIGS.  10 (A),  10 (B),  10 (C) , and  10 (D) are diagrams illustrating torque ripples in a case where the present embodiment is applied. 
         FIGS.  11 (A),  11 (B),  11 (C) , and  11 (D) are diagrams illustrating rotation orders of pulsation in a case where the present embodiment is applied. 
         FIGS.  12 (A),  12 (B),  12 (C) , and  12 (D) are views illustrating pulsation in a case where the present embodiment is applied. 
         FIG.  13    is a configuration diagram of an electric vehicle system according to the present embodiment. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
       FIG.  1    is an overall configuration diagram of a motor drive system including a motor control device  200 . 
     As illustrated in  FIG.  1   , the motor drive system includes a DC power source  100 , the motor control device  200 , and a motor  300 . The motor control device  200  converts DC power supplied from the DC power source  100  into AC power to drive the motor  300 . The DC power source  100  is mainly a secondary battery, and is a lithium ion battery or a nickel-metal hydride battery. 
     The motor control device  200  includes a first inverter circuit  201 , a second inverter circuit  202 , a smoothing capacitor  203 , a first current sensor  204 , a second current sensor  205 , a magnetic pole position sensor  206 , a magnetic pole position detector  207 , a control unit  208 , and a PWM signal drive circuit  209 . 
     The first inverter circuit  201  includes switching elements respectively corresponding to upper arms and lower arms of a U-phase, a V-phase, and a W-phase. The switching element includes an IGBT  221  and a diode  222 , and the upper arm and the lower arm are packaged together to form a power module  223 . The switching element may be a metal oxide semiconductor field effect transistor (MOSFET). The first inverter circuit  201  forms a three-phase bridge circuit by using three power modules  223 , and switches energization to each winding of a first-system winding set  301  of the motor  300 . The power module  223  may include a total of six switching elements of upper and lower arms for three phases in one package. 
     The second inverter circuit  202  has a redundant system inverter configuration provided in parallel with the first inverter circuit  201  with respect to the DC power source  100  and the smoothing capacitor  203 . Since the configuration of the second inverter circuit  202  is similar to that of the first inverter circuit  201 , the description thereof will be omitted. The second inverter circuit  202  forms a three-phase bridge circuit by using a power module, and switches energization to each winding of a second-system winding set  302  of the motor  300 . 
     The smoothing capacitor  203  suppresses pulsation of a voltage input from the DC power source  100  to the first inverter circuit  201  and the second inverter circuit  202  to perform smoothing. In the following description, the first inverter circuit  201  and the second inverter circuit  202  may be collectively referred to as inverter circuits  201  and  202 . In addition, a voltage detector  101  detects a DC voltage value of the DC power source  100  and outputs the detected value to the control unit  208 . 
     The first current sensor  204  is provided between an output line of the first inverter circuit  201  and the motor  300 . The second current sensor  205  is provided between an output line of the second inverter circuit  202  and the motor  300 . The first current sensor  204  detects first-system three-phase AC currents Iu 1 , Iv 3  , and Iw 1  (U-phase AC current Iu 1 , V-phase AC current Iv 1 , and W-phase AC current Iw 1 ) that energize the motor  300 . The second current sensor  205  detects second-system three-phase AC currents Iu 2 , Iv 2 , and Iw 2  (U-phase AC current Iu 2 , V-phase AC current Iv 2 , and W-phase AC current Iw 2 ) that energize the motor  300 . 
     The first current sensor  204  and the second current sensor  205  are configured by using, for example, a Hall current sensor or the like. Detection results of the two-system three-phase AC currents Iu 1 , Iv 1 , Iw 1 , Iu 2 , Iv 2 , and Iw 2  by the first current sensor  204  and the second current sensor  205  are input to the control unit  208  and used for generation of a gate signal, which is performed by the control unit  208 . In a double three-phase motor inverter, an example in which the first current sensor  204  and the second current sensor  205  are configured by three current sensors in each of the first system and the second system is shown. However, the number of current sensors may be set to two in each system, and an AC current of the remaining one phase may be calculated from the fact that the sum of the three-phase AC currents Iu, Iv, and Iw is zero. Further, a pulse-like DC current flowing from the DC power source  100  into the inverter circuits  201  and  202  is detected by a shunt resistor or the like inserted between the smoothing capacitor  203  and the inverter circuits  201  and  202 . Then, the two-system three-phase AC currents Iu 1 , Iv 1 , Iw 1 , Iu 2 , Iv 2 , and Iw 2  may be obtained based on the DC current and two-system three-phase AC voltages Vu 1 , Vv 1 , Vw 1 , Vu 2 , Vv 2 , and Vw 2  applied from the inverter circuits  201  and  202  to the motor  300 . 
     The magnetic pole position sensor  206  that detects the magnetic position θ is attached to the motor  300 . The magnetic pole position sensor  206  is more preferably a resolver including an iron core and a winding, and may be a sensor using a Hall element or a magnetoresistive element such as a GMR, sensor. 
     A signal from the magnetic pole position sensor  206  is input to the magnetic pole position detector  207 . The magnetic pole position detector  207  calculates the magnetic pole position θ from the input signal. The magnetic pole position detector  207  may estimate the magnetic position θ by using the two-system three-phase AC currents Iu 1 , Iv 1 , Iw 1 , Iu 2 , Iv 2 , and Iw 2  flowing through the motor  300  or the two-system three-phase AC voltages Vu 1 , Vv 1 , Vw 1 , Vu 2 , Vv 2 , and Vw 2  applied from the inverter circuits  201  and  202  to the motor  300  without using the input signal from the magnetic pole position sensor  206 . 
     The control unit  208  receives inputs of current values from the first current sensor  204  and the second current sensor  205  and the magnetic position e from the magnetic pole position detector  207 , and further receives an input of a torque command value corresponding to a target torque from a host controller or the like (not illustrated). The magnetic position θ is used in phase control of AC power, which is performed in a manner that the control unit  208  generates the gate signal in accordance with the phase of the induced voltage of the motor  300 . The control unit  208  performs PWM control based on input information to generate a PWM signal for driving the motor  300 , and outputs the PWM signal to the PWM signal drive circuit  209 . 
     The PWM signal drive circuit  209  generates a gate signal for controlling each of the switching elements included in the first inverter circuit  201  and the second inverter circuit  202  based on the PWM signal input from the control unit  208 . Then, the PWM signal drive circuit  209  outputs the gate signal to the inverter circuits  201  and  202 . 
     In the inverter circuits  201  and  202 , each of the switching elements is controlled in accordance with the gate signal input from the PWM signal drive circuit  209 , so that DC power supplied from the DC power source  100  is converted into AC power and output to the motor  300 . The smoothing capacitor  203  smooths the DC power supplied from the DC power source  100  to the inverter circuits  201  and  202 . 
     The motor  300  is a synchronous motor rotationally driven by AC power supplied from the inverter circuits  201  and  202 , and includes a stator and a rotor. The stator of the motor  300  is provided with two-system three-phase windings being the first-system winding set  301  and the second-system winding set  302 . AC power is input from the first inverter circuit  201  to the first-system winding set  301 , three-phase AC currents Iu 1 , Iv 1 , and Iw 1  flow through the respective windings forming the first-system winding set  301 , and an armature magnetic flux is generated in the respective windings. 
     Similarly, AC power is input from the second inverter circuit  202  to the second-system winding set  302 , three-phase AC currents Tu 2 , Iv 2 , and Iw 2  flow through the respective windings forming the second-system winding set  302 , and an armature magnetic flux is generated in the respective windings. Torque is generated in the rotor by generating an attraction force and a repulsive force between the combined magnetic flux of the armature magnetic flux generated in each of the windings of the two systems and the magnet magnetic flux of a permanent magnet arranged in the rotor. Thus, the rotor is rotationally driven. 
     Although  FIG.  1    illustrates one control unit  208  and one PWM signal drive circuit  209 , each of the inverter circuits  201  and  202  may include one control unit  208  including a PWM signal drive circuit. Furthermore, each of the inverter circuits  201  and  202  may have a PWM signal drive circuit  209  and a control unit  208 . 
     The control unit  208  receives the torque command value T* from the host controller or the like (not illustrated), and calculates current phases of currents to be energized by the respective inverter circuits  201  and  202  of the first system and the second system based on the torque command value T*. Furthermore, the control unit  208  calculates a voltage command value so that the current energized by the inverter circuits  201  and  202  of the first system and the second system has a desired current phase. Then, the control unit  208  generates a PWM signal based on the three-phase current command value of each of the first system and the second system, and outputs the PWM signal to the PWM signal drive circuit  209 . The PWM signal drive circuit  209  generates a gate signal based on the received PWM signal, and drives the switching elements of the inverter circuits  201  and  202 . 
     In addition, a storage unit  218  that stores various maps is connected to the control unit  208 . Although described in detail later, the control unit  208  shifts the phase of a PWM carrier signal used to generate a PWM signal for controlling the operation of each of the first inverter circuit  201  and the second inverter circuit  202  by using, a reference, the pulsation of an electromagnetic force caused by the magnetic circuit of the motor  300 . At this time, processing is performed with reference to a map stored in the storage unit  218  in advance. The control unit  208  is, for example, a microcomputer. The storage unit  218  may be provided in the control unit  208 . 
     Originally, it is ideal to drive the motor  300  with a sinusoidal current, but since the motor  300  that performs a variable speed operation needs to control the frequency of the current flowing from the inverter circuits  201  and  202  to the motor  300  in accordance with the rotational speed of the motor  300 , most of the motors  300  that perform the variable speed operation are driven by the inverter circuits  201  and  202 . 
     The PWM control performed by the control unit  208  is classified into two types by a difference in a control form of a frequency (carrier frequency) of the PWM carrier signal used to generate the PWM signal. Specifically, there are asynchronous PWM control in which the carrier frequency is constant regardless of the frequency of the current flowing through the motor  300 , and synchronous PWM control in which the carrier frequency is controlled to be an integer multiple of the frequency of the current flowing through the motor  300 . In a case where the motor  300  is driven to rotate at a high speed by using the asynchronous PWM control, the waveform of the current flowing through the motor  300  does not become a three-phase symmetrical waveform, which causes electromagnetic force pulsation of the motor  300 . In a case where the synchronous PWM control is used, the waveform of the current flowing through the motor  300  becomes a three-phase symmetrical waveform. Thus, it is possible to expect an effect of reducing the electromagnetic force pulsation of the motor  300  as compared with the asynchronous PWM control. 
     The pulsation of the electromagnetic force generated by the motor  300  is a change in the electromagnetic force generated in the rotor by applying a current from the inverter circuits  201  and  202  to the motor  300 . The pulsation of the electromagnetic force generated by the motor  300  is roughly divided into a torque ripple that is a pulsation component generated in a circumferential direction of the motor  300  and an electromagnetic excitation force that is a pulsation component generated in a radial direction of the motor  300 . Two main causes of the pulsation of the electromagnetic force in the motor  300  are a change in the electromagnetic force generated depending on the shape of the motor magnetic circuit including the core of the stator of the motor  300 , the coil of the stator, the core of the rotor, and the magnet of the rotor, and a change in the electromagnetic force generated by the harmonic wave included in the current applied from the inverter circuits  201  and  202  to the coil of the motor  300  due to the control of the inverter circuits  201  and  202 . 
     Since the high-rotation motor  300  generally performs weak field control, the magnitude and the phase of pulsation of the electromagnetic force caused by the magnetic circuit vary even with the same torque. Furthermore, the magnitude of the weak field depends also on the DC voltage of the DC power source  100 . The generation factor of the harmonic component included in the current applied from the inverter circuits  201  and  202  to the coil of the motor  300  is that the inverter circuits  201  and  202  are controlled by PWM control, and the voltage is applied not by a sine wave but by a PWM signal. The pulse amplitude of the PWM signal depends on the DC voltage. 
     Here, in the motor drive system in which the inverter circuits  201  and  202  of two systems are connected to the motor  300  including windings of two systems in which neutral points  303  and  304  are independent in a stator and the motor is driven, the pulsation of the electromagnetic force is determined by three factors of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  including the windings of two systems, the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201 , and the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202 . 
       FIG.  2    is a diagram illustrating, as an example, a torque ripple that is a circumferential component of the pulsation of the electromagnetic force of the motor  300  in a case where the present embodiment is not applied.  FIG.  2 (A)  is a diagram illustrating torque acting on the shaft of the motor  300 .  FIG.  2 (B)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 .  FIG.  2 (C)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force generated in the motor due to the control of the first inverter circuit  201 .  FIG.  2 (D)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force generated in the motor due to the control of the second inverter circuit  202 . The horizontal axis represents an electrical angle, and the vertical axis represents the torque.  FIGS.  2 (B),  2 (C) , and  2 (D) illustrate the torque ripple being the circumferential component of the pulsation of the electromagnetic force due to each factor in the motor  300 . 
     A torque ripple of the shaft of the motor  300  is generated as illustrated in  FIG.  2 (A)  by adding the circumferential component of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  illustrated in  FIG.  2 (B) , the circumferential component of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  illustrated in  FIG.  2 (C) , and the circumferential component of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  illustrated in  FIG.  2 (D) . This torque ripple becomes a vibration and noise of the motor  300 . 
       FIG.  3    is a diagram illustrating, as an example, a torque ripple that is a circumferential component of the pulsation of the electromagnetic force of the motor  300  in a case where the present embodiment is applied.  FIG.  3 (A)  is a diagram illustrating the torque of the shaft of the motor  300 .  FIG.  3 (B)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 .  FIG.  3 (C)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201 .  FIG.  3 (D)  is a diagram illustrating the circumferential component of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202 . The horizontal axis represents an electrical angle, and the vertical axis represents the torque. Similarly to  FIG.  2   ,  FIGS.  3 (B),  3 (C) , and  3 (D) illustrate the torque ripple being the circumferential component of the pulsation of the electromagnetic force due to each factor in the motor  300 . 
     In the present embodiment, the torque ripple finally generated by the motor  300  can be reduced by adjusting the phase of the pulsation of the electromagnetic force due to three factors by the control described later. Controllable elements in the pulsation of the electromagnetic force by the three elements are pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  and pulsation of the electromagnetic force caused by the control of the second inverter circuit  202 . The control unit  208  adjusts the phase e of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  and the phase θ 12  of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202 , by using, as a reference, the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . The phase θ I1  of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  is adjusted by adjusting the phase (carrier phase θ C1 ) of the PWM carrier signal used to generate the PWM signal for controlling the first inverter circuit  201 . The phase θ I2  of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  is adjusted by adjusting the phase (carrier phase θ C2 ) of the PWM carrier signal used to generate the PWM signal for controlling the second inverter circuit  202 . 
     As illustrated in  FIG.  3 (C) , for example, the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  is shifted by, for example, 20 degrees with respect to the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  illustrated in  FIG.  3 (B) . Furthermore, for example, the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  illustrated in  FIG.  3 (D)  is shifted by, for example, 40 degrees with respect to the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  illustrated in  FIG.  3 (B) . As a result, as illustrated in  FIG.  3 (A) , it is possible to suppress the pulsation generated in the torque of the shaft of the motor  300  and to suppress a vibration and noise of the motor  300 . 
     In the above description, it has been described that the torque ripple, which is the circumferential component of the electromagnetic force pulsation generated in the motor  300 , is reduced by adjusting the carrier phases θ C1  and θ C2  to adjust the phase θ I1  of the circumferential component in the electromagnetic force pulsation caused by the control of the first inverter circuit  201  and the phase θ I2  of the circumferential component in the electromagnetic force pulsation caused by the control of the second inverter circuit  202 . Here, the electromagnetic excitation force that is a radial component of the electromagnetic force pulsation generated in the motor  300  can be similarly controlled. That is, it is possible to reduce the electromagnetic excitation force that is the radial component of the electromagnetic force pulsation generated in the motor  300  by adjusting the carrier phases θ C1  and θ C2  to adjust the phase of the radial component in the electromagnetic force pulsation caused by the control of the first inverter circuit  201  and the phase of the radial component in the electromagnetic force pulsation caused by the control of the second inverter circuit  202 . 
       FIGS.  4 (A) and  4 (B)  are diagrams illustrating motor pulsation maps.  FIG.  4 (A)  is a pulsation map of the electromagnetic force for the circumferential component of the motor  300 , and  FIG.  4 (B)  is a pulsation map of the electromagnetic force for the radial component of the motor  300 . Both are stored in the storage unit  218  in advance. 
     As illustrated in  FIG.  4 (A) , the pulsation map of the electromagnetic force for the circumferential component of the motor  300  is a map in which current command values Id and Iq at the time of controlling the motor  300  are associated with a phase θ Tr  of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . This map is set corresponding to each of DC voltages Vdc 1 , Vdc 2 , and Vdc 3  of the DC power source  100 . For easy description,  FIG.  4 (A)  illustrates an example in which the motor pulsation map for the circumferential component is set for the three DC voltages Vdc 1 , Vdc 2 , and Vdc 3 . However, the motor pulsation map for the circumferential component only needs to be set corresponding to a plurality of DC voltages, and may be other than three. 
     As illustrated in  FIG.  4 (B) , the pulsation map of the electromagnetic force for the radial component of the motor  300  is a map in which the current command values Id and Iq at the time of controlling the motor  300  are associated with the phase θ Tr  of the pulsation caused by the magnetic circuit of the motor  300 . This map is set corresponding to each of DC voltages Vdc 1 , Vdc 2 , and Vdc 3  of the DC power source  100 . For easy description,  FIG.  4 (B)  illustrates an example in which the motor pulsation map for the radial component is set for the three DC voltages Vdc 1 , Vdc 2 , and Vdc 3 . However, the motor pulsation map for the radial component only needs to be set corresponding to a plurality of DC voltages, and may be other than three. 
     The motor pulsation maps illustrated in  FIGS.  4 (A) and  4 (B)  are stored in advance by using experimental values and design values. For example, in the case of the DC voltage Vdc 1  of the DC power source  100 , the motor drive system illustrated in  FIG.  1    is operated to obtain certain current command values Id and Iq and the phase θ Tr  of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 , and thus to obtain a map. Then, the phase θ Tr  of each electromagnetic force pulsation is obtained by variously changing the current command values Id and Iq to obtain a map. Similarly, the DC voltage of the DC power source  100  is changed to obtain a map. In this case, which of the circumferential component and the radial component of the electromagnetic force pulsation is to be reduced is selected in accordance with the rotational speed of the motor  300 . For example, in a case where the rotational speed of the motor  300  is low, the map for the circumferential component illustrated in  FIG.  4 (A)  is used. In a case where the rotational speed of the motor  300  is high, the map for the radial component illustrated in  FIG.  4 (B)  is used. In the motor pulsation map, the magnitude of pulsation of the electromagnetic force (torque ripple generated in the circumferential direction and electromagnetic excitation force generated in the radial direction) caused by the magnetic circuit of the motor  300  based on the current command values Id and Iq and a reference phase with respect to the electrical angle of the motor  300  are also stored in advance. 
     In a case where the control unit  208  refers to the map in the storage unit  218  and the rotational speed of the motor  300  is low, the phase θ Tr  of the electromagnetic force pulsation corresponding to the current command values Id and Iq is obtained with reference to the map for the circumferential component illustrated in  FIG.  4 (A) . In a case where the rotational speed of the motor  300  is high, the phase of the electromagnetic force pulsation corresponding to the current command values Id and Iq is obtained with reference to the map for the radial component illustrated in  FIG.  4 (B) . 
       FIGS.  5 (A) and  5 (B)  are diagrams illustrating the circumferential carrier phase map.  FIG.  5 (A)  is a carrier phase map for a first circumferential component of the first inverter circuit  201 , and  FIG.  5 (B)  is a carrier phase map for a second circumferential component of the second inverter circuit  202 . Both are stored in the storage unit  218  in advance. 
     As illustrated in  FIG.  5 (A) , the carrier phase map for the first circumferential component of the first inverter circuit  201  is a map in which the current command values Id and Iq at the time of controlling the motor  300  are associated with a carrier phase θ C1  for reducing the circumferential component of the electromagnetic force pulsation caused by the control of the first inverter circuit  201 . As illustrated in  FIG.  5 (B) , the carrier phase map for the second circumferential component of the second inverter circuit  202  is a map in which the current command values Id and Iq at the time of controlling the motor  300  are associated with a carrier phase for reducing the circumferential component of the electromagnetic force pulsation caused by the control of the second inverter circuit  202 . The map is set corresponding to each of DC voltages Vdc 1 , Vdc 2 , and Vdc 3  of the DC power source  100 . For convenience of description,  FIGS.  5 (A) and  5 (B)  illustrate an example in which the carrier phase map for the first circumferential component and the carrier phase map for the second circumferential component are respectively set for the three DC voltages Vdc 1 , Vdc 2 , and Vdc 3 , but the maps only needs to be set corresponding to a plurality of DC voltages and may be other than three. 
       FIGS.  6 (A) and  6 (B)  are diagrams illustrating the radial carrier phase map.  FIG.  6 (A)  is a carrier phase map for a first radial component of the first inverter circuit  201 , and  FIG.  6 (B)  is a carrier phase map for a second radial component of the second inverter circuit  202 . Both are stored in the storage unit  218  in advance. 
     As illustrated in  FIG.  6 (A) , the carrier phase map for the first radial component of the first inverter circuit  201  is a map in which the current command values Id and Iq at the time of controlling the motor  300  are associated with a carrier phase e for reducing the radial component of the electromagnetic force pulsation caused by the control of the first inverter circuit  201 . As illustrated in  FIG.  6 (B) , the carrier phase map for the second radial component of the second inverter circuit  202  is a map in which the current command values Id and Iq at the time of controlling the motor  300  are associated with a carrier phase θ C2  for reducing the radial component of the electromagnetic force pulsation caused by the control of the second inverter circuit  202 . The map is set corresponding to each of DC voltages Vdc 1 , Vdc 2 , and Vdc 3  of the DC power source  100 . For convenience of description,  FIGS.  6 (A) and  6 (B)  illustrate an example in which the carrier phase map for the first radial component and the carrier phase map for the second radial component are respectively set for the three DC voltages Vdc 1 , Vdc 2 , and Vdc 3 , but the maps only needs to be set corresponding to a plurality of DC voltages and may be other than three. 
     The carrier phases θ C1  and θ C2  in the circumferential carrier phase maps of  FIGS.  5 (A) and  5 (B)  and the radial carrier phase maps of  FIGS.  6 (A) and  6 (B)  are represented by using, a reference, the phase θ Tr  in the motor pulsation maps of  FIGS.  4 (A) and  4 (B) . That is, a phase difference between the electromagnetic force pulsation caused by the magnetic circuit of the motor  300  and each PWM carrier signal for reducing the electromagnetic force pulsation caused by the control of each of the first inverter circuit  201  and the second inverter circuit  202  is represented by each of the circumferential carrier phase maps of  FIGS.  5 (A) and  5 (B)  and the radial carrier phase maps of  FIGS.  6 (A) and  6 (B) . 
     The PWM carrier phase maps of the circumferential components illustrated in  FIGS.  5 (A) and  5 (B)  are stored in advance by using experimental values and design values. For example, in the case of the DC voltage Vdc 1  of the DC power source  100 , the motor drive system illustrated in  FIG.  1    is operated to obtain certain current command values Id and Iq and carrier phases θ C1  and θ C2  for reducing the circumferential component of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  and the second inverter circuit  202 , and to obtain a map. Specifically, the phase of the PWM carrier signal of the first inverter circuit  201  is shifted, and the phase when the torque ripple of the motor  300  becomes the smallest is obtained as the carrier phase θ C1 . The same applies to the second inverter circuit  202 . That is, if the phases of the PWM carrier signals used for the PWM control of the inverter circuits  201  and  202  are shifted to the phases θ C1  and θ C2 , the pulsation of the electromagnetic force in the circumferential direction caused by the control can be minimized. Then, the current command values Id and Iq are variously changed, and the carrier phases θ C1  and θ C2  in each case are obtained and used as a map. Similarly, the DC voltage of the DC power source  100  is changed to obtain a map. 
     Similarly, the PWM carrier phase maps of the radial components illustrated in  FIGS.  6 (A) and  6 (B)  are stored in advance by using experimental values and design values. For example, in the case of the DC voltage Vdc 1  of the DC power source  100 , the motor drive system illustrated in  FIG.  1    is operated to obtain certain current command values Id and Iq and carrier phases θ C1  and θ C2  for reducing the radial component of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  and the second inverter circuit  202 , and to obtain a map. Then, the current command values Id and Iq are variously changed, and the carrier phases θ C1  and θ C2  in each case are obtained and used as a map. Similarly, the DC voltage of the DC power source  100  is changed to obtain a map. 
     In a case where the rotational speed of the motor  300  is low, the map for the circumferential component illustrated in  FIGS.  5 (A) and  5 (B)  is used. In a case where the rotational speed of the motor  300  is high, the map for the radial component illustrated in  FIGS.  6 (A) and  6 (B)  is used. 
     In a case where the control unit  208  refers to the map in the storage unit  218  and the rotational speed of the motor  300  is low, the carrier phases θ C1  and θ C2  corresponding to the current command values Id and Iq are obtained with reference to the map for the circumferential component illustrated in  FIGS.  5 (A) and  5 (B) . In a case where the rotational speed of the motor  300  is high, the carrier phases θ C1  and θ C2  corresponding to the current command values Id and Iq is obtained with reference to the map for the radial component illustrated in  FIGS.  6 (A) and  6 (B) . 
     Next, the pulsation (electromagnetic excitation force) of the electromagnetic force in the radial direction caused by the magnetic circuit of the motor  300  will be described with reference to  FIG.  7   . 
     In a case where the motor  300  rotates at a high speed, the number of pulses/period decreases on the high speed rotation side. Thus, the synchronous PWM control is used in the present embodiment. In a case where the synchronous PWM control is used, the electromagnetic force pulsation in the radial direction caused by the magnetic circuit of the motor  300  and the electromagnetic force pulsation in the radial direction caused by the harmonic current caused by the control of the inverter circuits  201  and  202  can be superimposed regardless of the rotational speed of the motor  300 . 
       FIG.  7    is a diagram illustrating a relationship between the rotation speed of the motor  300  and the excitation frequency (frequency of pulsation in the radial direction). The horizontal axis represents the rotation speed of the motor  300 , and the vertical axis represents the excitation frequency. 
     The asynchronous PWM control is used until the rotation speed of the motor  300  reaches 12000 rpm, and the synchronous PWM control is used in a case where the rotation speed of the motor  300  exceeds 12000 rpm. 
       FIG.  7    illustrates an example in which the motor  300  is an 8-pole motor (the number of poles P=8). In the example of  FIG.  7   , the carrier frequency fc is 10 kHz in the asynchronous PWM control. In the synchronous PWM control, the carrier frequency fc is adjusted to form a PWM carrier signal of nine pulses for each cycle of the voltage command. 
     First, the excitation frequency caused by the magnetic circuit of the motor  300  will be described. An electrical angle (fundamental harmonic current) for turning the motor  300  is defined as a frequency f 1  [Hz]. A relationship between the rotation speed N [rpm] of the motor  300  and the electrical angular frequency f 1  is represented by the following Expression (1). P is the number of poles of the motor  300 . 
         f 1= N/ 60× P/ 2[rpm]  (1)
 
     An excitation frequency f 6  of the sixth order of rotation (electrical angle) of the motor  300  is represented by the following Expression (2). An excitation frequency f 12  of the 12th order of the rotation (electrical angle) of the motor  300  is represented by the following Expression (3). 
         f 6=6× f 1[Hz]  (2)
 
         f 12=12× f 1[Hz]  (3)
 
     In  FIG.  7   , f 6  illustrated is the excitation frequency f 6  of the sixth order of rotation (electrical angle), and f 12  illustrated is the excitation frequency f 12  of the 12th order of rotation (electrical angle). As illustrated in  FIG.  7   , the excitation frequencies f 6  and f 12  linearly increase from a region of the asynchronous PWM control to a region of the synchronous PWM control. A carrier frequency fc is constant in the asynchronous PWM control. 
     Next, the excitation frequency (frequency of pulsation in the radial direction) by the harmonic current caused by the inverter circuits  201  and  202  will be described. The carrier frequency fc and the sideband component of fc±3f 1  become the excitation frequency of the annular 0th order. The annular 0th order is the rotation order of pulsation in the radial direction of the motor  300 . The component that is a radial component of the electromagnetic force generated in a gap of the motor  300  and uniformly changes with time in the radial direction is referred to as an annular 0th order mode. In the present embodiment, the pulsation of the electromagnetic force of the motor  300  in the radial direction is reduced for the annular 0th order radial pulsation. 
     In the case of the asynchronous PWM control, for example, when the rotation speed of the motor  300  is 6000 rpm, fc±3f1 is represented by the following Expressions (4) and (5). 
         fc+ 3 f 1=10000+3×6000/60×8/2=11200[Hz]  (4)
 
         fc− 3 f 1=10000−3×6000/60×8/2=8800[Hz]  (5)
 
     In the case of the synchronous PWM control, assuming the PWM carrier signal of nine pulses for each cycle of the voltage command, the carrier frequency fc is represented by the following Expression (6). Therefore, the sideband components are represented by the following Expressions (7) and (8), respectively. 
         fc= 9× f 1 [Hz]  (6)
 
         fc+ 3 f 1=9× f 1+3× f 1=12× f 1[Hz]  (7)
 
         fc− 3 f 1=9× f 1−3× f 1=6&#39; f 1[Hz]  (8)
 
     As illustrated in  FIG.  7   , in the case of the synchronous PWM control, the frequencies f 6  and f 12  of the pulsation of the electromagnetic force in the radial direction caused by the magnetic circuit of the motor  300  overlap with the frequencies fc+3f 1  and fc-3f 1  of the pulsation of the electromagnetic force in the radial direction caused by the control of the inverter circuits  201  and  202 . As described above, since the electromagnetic force pulsation in the radial direction caused by the magnetic circuit of the motor  300  and the electromagnetic force pulsation in the radial direction caused by the control of the inverter circuits  201  and  202  have the same frequency, it is possible to cancel the electromagnetic excitation force caused by the magnetic circuit of the motor  300  by shifting the phase of the electromagnetic force pulsation in the radial direction caused by the control of the inverter circuits  201  and  202  by the control to be described later. 
       FIG.  8    is a diagram illustrating the frequency of the voltage command and the carrier frequency fc.  FIG.  8 (A)  illustrates a waveform by the first inverter circuit  201 , and  FIG.  8 (B)  illustrates a waveform by the second inverter circuit  202 . The left of each drawing illustrates a case where the rotational speed of the motor  300  is low, and the right of each drawing illustrates a case where the rotational speed of the motor  300  is high. 
     The control of the control unit  208  illustrated in  FIG.  1    is synchronous PWM control, and the control unit  208  controls the frequency of the voltage command and the carrier frequency fc. The frequency of the voltage command is a frequency f 1 [Hz] of an electrical angle (fundamental harmonic current) for turning the motor  300 . As illustrated on the left of  FIGS.  8 (A) and  8 (B) , in the low speed rotation, the carrier frequency fc is controlled to form a PWM carrier signal of nine pulses for each cycle of the frequency f 1  of the voltage command. As illustrated on the right side of  FIGS.  8 (A) and  8 (B) , the carrier frequency fc is similarly controlled to form a PWM carrier signal of nine pulses for each cycle of the frequency f 1  of the voltage command even in the high speed rotation. The waveform by the first inverter circuit  201  and the waveform by the second inverter circuit  202  are similar waveforms. In this example, an example in which the PWM carrier signal of nine pulses is formed for each cycle of the frequency f 1  of the voltage command has been described, but the frequency of the PWM carrier signal only needs to be an integer multiple of the frequency of the voltage command. In particular, it is preferable to perform control such that the integer multiple is an odd-numbered integer multiple or an integer multiple of a multiple of 3. 
     That is, the control unit  208  adjusts the carrier frequency fc such that the frequency of the PWM carrier signal used in the PWM control of each of the first inverter circuit  201  and the second inverter circuit  202  becomes an integer multiple of the frequency of the voltage command for driving the motor  300  in synchronization with the frequency f 1  of the voltage command for driving the motor  300 . By adjusting the carrier frequency in this manner, in a case where the synchronous PWM control is used, it is possible to superimpose the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  and the pulsation of the electromagnetic force caused by the control of the inverter circuits  201  and  202 , regardless of the rotational speed of the motor  300 . Since both pulsations have the same frequency, it is possible to cancel the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  by shifting the phase of the pulsation of the electromagnetic force caused by the control of the inverter circuits  201  and  202  by the control to be described later. 
       FIG.  9    is a flowchart illustrating processing of the control unit  208  in the motor control device  200 . 
     The flowchart illustrated in  FIG.  9    is performed at regular time intervals or every time the torque command value T* is input. A program illustrated in this flowchart can be executed by a computer including a CPU, a memory, and the like. All or a portion of the processing may be realized by a hard logic circuit. Furthermore, the program can be provided by being stored in advance in a storage medium of the motor control device  200 . Alternatively, the program may be stored and provided in an independent storage medium, or the program may be recorded and stored in the storage medium of the motor control device  200  via a network line. Various forms of computer-readable computer program products, such as data signals (carrier waves), may be provided. 
     In Step S 901  of  FIG.  9   , the control unit  208  receives a torque command value T* from the host controller or the like. Then, in Step S 902 , the control unit  208  creates current command values Id and Iq from the received torque command value T*. 
     Then, in Step S 903 , as described with reference to  FIGS.  8 (A) and  8 (B) , the control unit  208  adjusts the frequency of a PWM carrier signal to be an integer multiple of the frequency of a voltage command. At this time, for example, it is desirable to adjust the integer multiple to an odd-numbered integer multiple or an integer multiple of a multiple of 3. 
     Then, in Step S 904 , the control unit  208  selects a pulsation component to be reduced. That is, the circumferential component is selected in a case where the rotational speed of the motor  300  is lower than a predetermined value, and the radial component is selected in a case where the rotational speed of the motor  300  is equal to or greater than the predetermined value. The value of the rotational speed of the motor  300  is determined based on the rotation position B from the magnetic pole position detector  207 . In other words, the control unit  208  selects one of a torque ripple generated in the circumferential direction in the electromagnetic force pulsation caused by the magnetic circuit of the motor  300  and the electromagnetic excitation force generated in the radial direction in the electromagnetic force pulsation caused by the magnetic circuit of the motor  300 , based on the rotational speed of the motor  300 . 
     Then, in Step S 905 , the control unit  208  searches a motor pulsation map stored in the storage unit  218 . As described above, the motor pulsation map includes the map for the circumferential component illustrated in  FIG.  4 (A)  and the map for the radial component illustrated in  FIG.  4 (B) . Since the pulsation component to be reduced is selected in Step S 904 , the map corresponding to the selected pulsation component is searched for. Prior to the search, the control unit  208  causes the voltage detector  101  to detect a DC voltage value of the DC power source  100 . That is, in a case where the rotational speed is lower than the predetermined value, the map corresponding to the detected DC voltage value of the DC power source  100  among the maps for the three circumferential components illustrated in  FIG.  4 (A)  is searched based on the current command values Id and Iq, and the phase θ Tr  of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  is acquired. In a case where the rotational speed is equal to or greater than the predetermined value, the map corresponding to the detected DC voltage value of the DC power source  100  among the maps for the three radial components illustrated in  FIG.  4 (B)  is searched based on the current command values Id and Iq, and the phase θ Tr  of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  is acquired. In general, the amplitude of the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  changes due to a change in the DC voltage of the DC power source  100 . In the present embodiment, the map corresponding to the DC voltage value detected by the voltage detector  101  is referred to among the motor pulsation maps set in advance for the plurality of DC voltage values, and thus it is also possible to handle a change in amplitude. 
     In Step S 906 , the control unit  208  estimates the phase of the pulsation caused by the magnetic circuit of the motor  300  from the phase θ Tr  searched in Step S 905 . Description will be made below with reference to  FIG.  10   . 
       FIG.  10    is a diagram illustrating a torque ripple in a case where the present embodiment is applied.  FIG.  10 (A)  is a diagram illustrating the magnetic position of the motor  300 . The horizontal axis represents time and the vertical axis represents an electrical angle.  FIG.  10 (B)  illustrates the torque of the shaft of the motor  300 . The horizontal axis represents time and the vertical axis represents torque.  FIG.  10 (C)  is a diagram illustrating a PWM carrier signal and a voltage command of the first inverter circuit  201 .  FIG.  10 (D)  is a diagram illustrating a PWM carrier signal and a voltage command of the second inverter circuit  202 . The horizontal axis represents time, and the vertical axis represents a voltage. 
     As illustrated in  FIG.  10 (A) , the magnetic position of the motor  300  changes at every electrical angle of  360  degrees with the rotation of the motor  300 , and the rotation angle of 0 degree is a reference position. As illustrated in  FIG.  10 (B) , the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  appearing in the torque of the shaft of the motor  300  is generated at a frequency of 6n times (n =6, 12, 18, per electrical angle cycle in the three-phase motor  300 . Since the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  is determined by the current command values Id and Iq, it is necessary to check the shift amount from the reference position of the motor  300 . 
     The control unit  208  estimates the phase of the pulsation caused by the magnetic circuit of the motor  300  by using the detection signal of the magnetic pole position detector  207  attached to the motor  300  and the current command value to the motor  300 . Since the pulsation of the electromagnetic force caused by the magnetic circuit with respect to the rotation angle of the motor  300  can be estimated, it is possible to adjust the carrier phases θ C1  and θ C2  as illustrated in  FIGS.  10 (C) and  10 (D)  by using this estimation as a reference as described later. 
     The description returns to the flowchart illustrated in  FIG.  9   . 
     In Step S 907  of  FIG.  9   , the control unit  208  searches a carrier phase map stored in the storage unit  218 . As described above, the carrier phase map includes the circumferential carrier phase map illustrated in  FIGS.  5 (A) and  5 (B)  and the radial carrier phase map illustrated in  FIGS.  6 (A) and  6 (B) . 
     Since the pulsation component to be reduced is selected in Step S 904 , the map corresponding to the selected pulsation component is searched for. Prior to the search, the control unit  208  causes the voltage detector  101  to detect a DC voltage value of the DC power source  100 . That is, in a case where the rotational speed is lower than the predetermined value, the map corresponding to the detected DC voltage value of the DC power source  100  among the maps for the three circumferential components illustrated in  FIGS.  5 (A) and  5 (B)  is searched based on the current command values Id and Iq, and the carrier phases θ C1 and θ C2  are acquired. In a case where the rotational speed is equal to or greater than the predetermined value, the map corresponding to the detected DC voltage value of the DC power source  100  among the maps for the three radial components illustrated in  FIGS.  6 (A) and  6 (B)  is searched based on the current command values Id and Iq, and the carrier phases θ C1  and θ C2  are acquired. 
     In general, the amplitude of the pulsation of the electromagnetic force caused by the control of the inverter circuits  201  and  202  changes due to a change in the DC voltage of the DC power source  100 . In the present embodiment, even though the amplitude of the pulsation of the electromagnetic force caused by the control of the inverter circuits  201  and  202  changes, it is possible to secure the effect of reducing the torque ripple and the electromagnetic excitation force when superimposed on the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 , by adjusting the phases. That is, the shift amount of the phase of the PWM carrier signal is adjusted based on the DC voltage value applied to the first inverter circuit  201  and the second inverter circuit  202  as described later. 
     In Step S 908  of  FIG.  9   , the control unit  208  shifts the PWM carrier signal for the first inverter circuit  201  by the phase θ C1  by using, as a reference, the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . Furthermore, the PWM carrier signal for the second inverter circuit  202  is shifted by the phase θ C2  by using, as a reference, the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . 
     In Step S 909 , the control unit  208  drives the first inverter circuit  201  and the second inverter circuit  202  to output an AC voltage to the motor  300 . 
     In this manner, the phase of the PWM carrier signal is shifted with reference to the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  by the combined wave of the harmonic currents energized by the first inverter circuit and the second inverter circuit. As a result, it is possible to suppress pulsation caused by the magnetic circuit of the motor  300 . 
       FIG.  3    (C) illustrates an example in which the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  is shifted by 20 degrees.  FIG.  3 (D)  illustrates an example in which the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  is shifted by 40 degrees. As described above, the phase θ I1  of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  is adjusted by adjusting the carrier phase θ C1  of the first inverter circuit  201 . The phase  812  of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  is adjusted by adjusting the carrier phase θ C2  of the second inverter circuit  202 . As a result, as illustrated in  FIG.  3 (A) , it is possible to suppress the torque ripple generated in an axial direction of the motor  300  and the electromagnetic excitation force generated in the radial direction of the motor  300 , and to suppress a vibration and noise of the motor  300 . 
     In the present embodiment, the motor pulsation map illustrated in  FIGS.  4 (A) and  4 (B) , the circumferential carrier phase map illustrated in  FIGS.  5 (A) and  5 (B) , and the radial carrier phase map illustrated in  FIGS.  6 (A) and  6 (B)  are selected and used in accordance with the rotational speed of the motor  300 . That is, one of the torque ripple generated in the circumferential direction of the pulsation caused by the magnetic circuit of the motor  300  and the electromagnetic excitation force generated in the radial direction of the pulsation caused by the magnetic circuit of the motor  300  is selected based on the rotational speed of the motor  300 , and the phase of the PWM carrier signal is shifted to reduce the selected torque ripple or electromagnetic excitation force. As a result, which one of the torque ripple generated in the circumferential direction and the electromagnetic excitation force generated in the radial direction becomes the cause of the vibration is changed depending on the rotational speed of the motor  300 , but it is possible to reduce the pulsation having the larger influence and to reduce the vibration. 
     In the present embodiment, the motor pulsation map illustrated in  FIGS.  4 (A) and  4 (B) , the circumferential carrier phase map illustrated in  FIGS.  5 (A) and  5 (B) , and the radial carrier phase map illustrated in  FIGS.  6 (A) and  6 (B)  are selected and used in accordance with the DC voltage value of the DC power source  100  detected by the voltage detector  101 . That is, the shift amount of the phase of the PWM carrier signal is adjusted based on the DC voltage applied to the first inverter circuit  201  and the second inverter circuit  202 . As a result, the amplitude of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  and the second inverter circuit  202  changes due to the change in the DC voltage of the DC power source  100 , but even though the amplitudes change, it is possible to secure the effect of reducing the torque ripple and the electromagnetic excitation force when superimposed on the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . 
     Next, the rotation order of the pulsation will be described with reference to  FIG.  11   . 
       FIG.  11 (A)  is a diagram illustrating the magnetic position of the motor  300 . The horizontal axis represents time and the vertical axis represents an electrical angle.  FIG.  11 (B)  illustrates the torque ripple of the shaft of the motor  300 . The horizontal axis represents time and the vertical axis represents torque.  FIG.  11 (C)  is a diagram illustrating an electrical angle sixth-order component of the torque ripple.  FIG.  11 (D)  is a diagram illustrating an electrical angle 12th-order component of the torque ripple. The horizontal axis represents time, and the vertical axis represents torque. 
     The torque ripple of the shaft of the motor  300  illustrated in  FIG.  11 (B)  indicates the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . By referring to the magnetic pole position detector  403 , it is possible to check how many degrees the rotor of the motor  300  is at in terms of electrical angle. The pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300  illustrated in  FIG.  11 (B)  is cut out by one electrical angle cycle (360 degrees) illustrated in  FIG.  11 (A)  and component analysis is performed, and then results as illustrated in  FIGS.  11 (C) and  11 (D)  are obtained. That is, in the waveform of  FIG.  11 (C) , there are six pulsations of the electromagnetic force in one electrical angle cycle (360 degrees). The pulsations occur 6 times with respect to one electrical angle rotation, which is referred to as an electrical angle sixth-order component. In the waveform of  FIG.  11 (D) , there are 12 pulsations of the electromagnetic force in one electrical angle cycle (360 degrees), and the pulsations occur 12 times with respect to one electrical angle rotation, which is referred to as an electrical angle 12th-order component. 
     Next, control for reducing the pulsation of the electromagnetic force of the electrical angle sixth-order component will be described.  FIG.  12    is a diagram illustrating the pulsation in a case where the present embodiment is applied.  FIG.  12 (A)  is a diagram illustrating the torque of the shaft of the motor  300 .  FIG.  12 (B)  is a diagram illustrating the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 .  FIG.  12 (C)  is a diagram illustrating the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201 .  FIG.  12 (D)  is a diagram illustrating the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202 . The horizontal axis represents an electrical angle, and the vertical axis represents the torque. 
     In the control for reducing the pulsation of the electromagnetic force of the electrical angle sixth-order component, the same processes as Steps S 901  to S 907  and S 909  described with reference to  FIG.  9    are executed, but the following processes are executed in Step S 908  of  FIG.  9   . 
     As illustrated in  FIG.  12 (C) , the control unit  208  shifts the phase θ I1  of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  by 30 degrees by using, as a reference, the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . Furthermore, as illustrated in  FIG.  12 (D) , the control unit  208  shifts the phase θ I2  of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  by 30 degrees by using, as a reference, the pulsation of the electromagnetic force due to the magnetic circuit of the motor  300 . 
     As a result, as illustrated in  FIG.  12 (A) , it is possible to reduce the electrical angle sixth-order component of the pulsation generated in the torque of the shaft of the motor  300  and to suppress a vibration and noise of the motor  300 . 
     Next, control for reducing the pulsation of the electromagnetic forces of the electrical angle sixth-order component and the electrical angle 12th-order component will be described. 
     As described above,  FIG.  3    is a diagram illustrating the pulsation of the electromagnetic force of the motor  300  in a case where the present embodiment is applied. 
     In the control for reducing the pulsation of the electromagnetic force of the electrical angle sixth-order component and the electrical angle 12th-order component, the same processes as Steps S 901  to S 907  and S 909  described with reference to  FIG.  9    are executed, but the following processes are executed in Step S 908  of  FIG.  9   . 
     As illustrated in  FIG.  3 (C) , the control unit  208  shifts the phase e of the pulsation of the electromagnetic force caused by the control of the first inverter circuit  201  by 20 degrees by using, as a reference, the pulsation of the electromagnetic force caused by the magnetic circuit of the motor  300 . Furthermore, as illustrated in  FIG.  3 (D) , the control unit  208  shifts the phase θ I2  of the pulsation of the electromagnetic force caused by the control of the second inverter circuit  202  by 40 degrees by using, as a reference, the pulsation of the electromagnetic force due to the magnetic circuit of the motor  300 . 
     As a result, as illustrated in  FIG.  3 (A) , it is possible to reduce the electrical angle sixth-order component and the electrical angle 12th-order component of the pulsation generated in the torque of the shaft of the motor  300  and to suppress a vibration and noise of the motor  300 . 
       FIG.  13    is a configuration diagram of an electric vehicle system according to the present embodiment. As illustrated in  FIG.  13   , the electric vehicle system includes a power train in which the motor  300  is applied as a motor/generator, and travels by using a rotational driving force of the motor  300 . The electric vehicle system will be described by using a hybrid system as an example. 
     In  FIG.  13   , a front wheel axle  801  is rotatably supported on a front portion of the electric vehicle  800 , and front wheels  802  and  803  are provided at both ends of the front wheel axle  801 . A rear wheel axle  804  is rotatably supported on a rear portion of the electric vehicle  800 , and rear wheels  805  and  806  are provided at both ends of the rear wheel axle  804 . 
     A differential gear  811  which is a power distribution mechanism is provided at a central portion of the front wheel axle  801 , and a rotational driving force transmitted from an engine  810  via a transmission  812  is distributed to the left and right front wheel axles  801 . 
     In the engine  810  and the motor  300 , a pulley provided on a crankshaft of the engine  810  and a pulley provided on a rotation shaft of the motor  300  are mechanically joined via a belt. As a result, the rotational driving force of the motor  300  can be transmitted to the engine  810 , and the rotational driving force of the engine  810  can be transmitted to the motor  300 . In the motor  300 , the three-phase AC power controlled by the motor control device  200  incorporating the inverter circuits  201  and  202  is supplied to the coil of the stator, so that the rotor rotates and generates a rotational driving force in corresponding to the three-phase AC power. The motor control device  200  is the device described above in the present embodiment. 
     That is, while the motor  300  is controlled by the motor control device  200  to operate as an electric motor, an electromotive force is induced in the coil of the stator by the rotation of the rotor by receiving the rotational driving force of the engine  810 , and thus the motor operates as a generator that generates three-phase AC power. 
     The motor control device  200  is a power conversion device that converts DC power supplied from the DC power source  100  that is a high-voltage battery into three-phase AC power, and controls a three-phase AC current flowing through a stator coil of the motor  300 , which corresponds to the magnetic position in accordance with an operation command value. 
     The three-phase AC power generated by the motor  300  is converted into DC power by the motor control device  200  to charge the DC power source  100 . The DC power source  100  is electrically connected to a low-voltage battery  823  via a DC-DC converter  824 . The low-voltage battery  823  constitutes a low-voltage (14 V) system power source of the electric vehicle  800 , and is used as a power source of a starter  825  that initially starts (cold-starts) the engine  810 , a radio, a light, or the like. 
     In general, regarding the vibration and the noise of the motor  300 , the vibration noise is generated in a manner that an excitation force generated by an electromagnetic force is transmitted through the main body of the motor  300  and the attached structure and shakes each portion. In addition, in a case where the natural mode and the frequency of the structure overlap with the excitation mode of the excitation force, a resonance state occurs, and the vibration noise is amplified. In the present embodiment, it is possible to reduce a vibration and noise of the motor  300  and to further reduce a vibration and noise of the electric vehicle  800  on which the motor  300  is mounted. 
     According to the embodiment described above, the following operational effects can be obtained. 
     (1) A motor control device  200  includes a first inverter circuit  201  and a second inverter circuit  202  of a redundant system, the first inverter circuit  201  and the second inverter circuit  202  controlling a motor  300 , and a control unit  208  that controls the first inverter circuit  201  and the second inverter circuit  202 . The first inverter circuit  201  converts the DC power into the AC power based on a PWM signal generated by using a first carrier signal. The second inverter circuit  202  converts the DC power into the AC power based on a PWM signal generated by using a second carrier signal. The control unit  208  shifts phases of the first carrier signal and the second carrier signal by using, as a reference, pulsation of an electromagnetic force caused by a magnetic circuit of the motor  300 . Thus, it is possible to suppress a vibration and noise generated in a motor. 
     (2) There is provided a motor control method in a motor control device  200  including a first inverter circuit  201  and a second inverter circuit  202  of a redundant system, the first inverter circuit  201  and the second inverter circuit  202  controlling a motor, and a control unit  208  that controls the first inverter circuit  201  and the second inverter circuit  202 . The motor control method includes converting, by the first inverter circuit  201 , the DC power into the AC power based on a PWM signal generated by using a first carrier signal, converting, by the second inverter circuit  202 , the DC power into the AC power based on a PWM signal generated by using a second carrier signal, and shifting, by the control unit, the phases of the first carrier signal and the second carrier signal by using, as a reference, pulsation of an electromagnetic force caused by a magnetic circuit of the motor  300 . Thus, it is possible to suppress a vibration and noise generated in a motor. 
     The present invention is not limited to the above-described embodiments, and other forms conceivable within the scope of the technical idea of the present invention are also included in the scope of the present invention as long as the characteristics of the present invention are not impaired. 
     REFERENCE SIGNS LIST 
       100  DC power source 
       101  voltage detector 
       200  motor control device 
       201  first inverter circuit 
       202  second inverter circuit 
       203  smoothing capacitor 
       204  first current sensor 
       205  second current sensor 
       206  magnetic pole position sensor 
       207  magnetic pole position detector 
       208  control unit 
       209  PWM signal drive circuit 
       223  power module 
       300  motor 
       301  first-system winding set 
       302  second-system winding set