Patent Publication Number: US-2018044116-A1

Title: Vibrating feeder control apparatus and vibrating feeder

Description:
TECHNICAL FIELD 
     The present invention relates to a vibrating feeder control apparatus for operating a vibrating feeder main body having an electromagnet as a driving source, and a vibrating feeder including the vibrating feeder control apparatus. 
     BACKGROUND ART 
     As a conventional vibrating feeder for conveying a work by utilizing vibration, there has been known a so-called linear feeder, which linearly vibrates a movable body and linearly conveys a work on a movable body (refer to Patent Literature 1), and a so-called bowl feeder, which conveys a work along the inner wall of a bowl by generating torsional vibration by twisting a bowl as a movable body to put a work in (refer to Patent Literature 2). 
     Although they are different in a vibrating direction, a movable body is elastically supported on a base to be easily displaced in a specific direction, and a driving force is given to the movable body side, whereby the above-mentioned linear vibration and torsional vibration can be generated in the movable body. 
     An electromagnet is often used as a driving source because of its low and easy control. By performing ON/OFF control of a current flowing in an electromagnet, a desired vibration can be generated in the movable body. However, harmonics and chatter often occur only by applying a simple pulse voltage to the electromagnet, and it is preferable to apply a sinusoidal AC voltage to perform quiet and smooth control. Further, in order to drive at a desired frequency, there is another way to generate a pseudo AC voltage using a PWM (Pulse Width Modulation) circuit, and supply the generated voltage to the electromagnet. 
     Further, for obtaining a large displacement, while reducing the energy supplied to the electromagnet, as well as applying an AC voltage of a frequency close to a resonance frequency of the vibrating feeder main body to the electromagnet, a so-called resonance point follow-up control may be performed, which causes the frequency of AC voltage to follow the resonance frequency changing constantly according to the weight and position of a work to be conveyed (see Patent Literature 2). 
     CITATION LIST 
     Patent literature 
     Patent Literature 1: Japanese Unexamined Patent Application Publication No. 3-106711 
     Patent Literature 2: Japanese Patent No. 4066480 
     SUMMARY OF THE INVENTION 
     Problems to be Solved by the Invention 
     However, in the conventional vibrating feeder including the above-described Patent Literature 2, for performing the resonance point follow-up control, it is necessary to provide a displacement sensor for detecting a displacement of the movable body on the vibrating feeder main body side. The control apparatus for driving the vibrating feeder main body is configured to control a drive frequency by determining whether a resonance state is established based on an output from the displacement sensor. 
     That is, for performing such a resonance point follow-up control, it is assumed that the displacement sensor is provided on the vibrating feeder main body side, which leads to an increase in the size and manufacturing cost of an apparatus to incorporate the displacement sensor. In addition, since it is necessary to connect the displacement sensor and the control apparatus with a cable, it takes time and labor for the wiring, and it is also necessary to take into account the possibility of failure due to disconnection or the likes. 
     Furthermore, for adding the function of the resonance point follow-up control to the existing vibrating feeder main body having no displacement sensor and appropriately control a movable body, it is necessary not only to update the control apparatus but also to add a displacement sensor to the vibrating feeder, requiring a large cost. 
     It is an object of the present invention to effectively solve such a problem, and in particular, to provide a vibrating feeder control apparatus of simple wiring and high reliability, which can control a drive frequency at which a phase difference between a voltage and a displacement becomes a predetermined relationship even when no displacement sensor is provided on the vibrating feeder main body side, and an inexpensive vibrating feeder using this control apparatus. 
     Solution to Problem 
     The present invention takes the following measures in order to achieve the above object. 
     That is, a vibrating feeder control apparatus of the present invention is an apparatus used for driving a vibrating feeder main body comprising a base, a movable body elastically supported by the base, an electromagnet provided on one of the base and the movable body, and a magnetic core provided on the other of the base and the movable body to oppose the electromagnet. The control apparatus comprises a PWM signal generation unit that generates a PWM signal based on a set drive frequency and applies a pseudo AC voltage corresponding to the PWM signal to the electromagnet, a current detection unit that detects a current flowing in the electromagnet by the pseudo AC voltage, a current change rate generation unit that generates a current change rate at a predetermined reference phase angle within one cycle of the pseudo AC voltage based on a detection value of the current detection unit, and a frequency correction unit that corrects the drive frequency based on a current change rate at a reference phase angle obtained by the current change rate generation unit. 
     With this configuration, by applying the pseudo AC voltage corresponding to the PWM signal generated by the PWM signal generation unit to the electromagnet, a pulse-like constant voltage is applied to the electromagnet in a short time. When a constant voltage is applied to the electromagnet in this way, the current change rate that is a gradient of the current flowing through the electromagnet corresponds to an inductance of the electromagnet. Since this inductance corresponds to a gap between the electromagnet and the magnetic core, in other words, a displacement amount of the movable body, obtaining the current change rate is to be regarded as knowing the displacement amount of the movable body at that point in time. Therefore, the current change rate generation unit generates a current change rate at a predetermined reference phase angle, and the frequency correction unit corrects a drive frequency based on the current change rate, whereby even when the displacement sensor for detecting a displacement of the movable body is not provided in the vibrating feeder main body, it is possible to control the drive frequency so that the phase difference between the voltage and the displacement becomes a predetermined relationship, and to realize a vibrating feeder with simple wiring, high reliability and low manufacturing cost. 
     To enable control of a drive frequency for more properly driving the vibrating feeder, it is preferable that a first reference phase angle and a second reference phase angle are set as the reference phase angle at positions that are substantially symmetrical around a phase angle at which a peak value of the pseudo AC voltage occurs, a first current change rate generation unit that generates a first current change rate corresponding to the first reference phase angle, and a second current change rate generation unit that generates a second current change rate corresponding to the second reference phase angle are provided as the current change rate generation unit, and the frequency correction unit is configured to correct the drive frequency based on the first and second current change rates obtained from the first and second current change rate generation units. 
     Further, for generating a current change rate easily and accurately when a constant voltage is being applied, it is preferable to configure the current change rate generation unit to generate the current change rate during a period from turn-on to turn-off of the PWM signal for one pulse corresponding to the first and second reference phase angles. 
     For obtaining a current change rate more accurately by avoiding the influence of a response delay immediately after switching to turn on the PWM signal, it is preferable to configure the current change rate generation unit to generate the current change rate during a period from turn-on of the PWM signal for one pulse corresponding to the first and second reference phase angles to turn-off after a lapse of predetermined time. 
     For easily realizing the resonance point follow-up control, it is preferable that the first reference phase angle is set to a range over −90° and below 0° and the second reference phase angle is set to a range over 0° and below 90° when a phase angle at which a peak value of the pseudo AC voltage occurs is assumed to be 0°, and the frequency correction unit does not correct a drive frequency when a current change rate difference obtained by subtracting an absolute value of the second current change rate from an absolute value of the first current change rate is within a predetermined range including a zero therebetween, and corrects the drive frequency in a decreasing direction when the current change rate difference exceeds the predetermined range, and corrects the drive frequency in an increasing direction when the current change rate difference is smaller than the predetermined range. 
     For realizing an inexpensive vibrating feeder with simple wiring and high reliability, it is preferable to configure to include the vibrating feeder control apparatus described in any one of the above and a vibrating feeder main body controlled by the vibrating feeder control apparatus. 
     Effect of the Invention 
     According to the present invention described above, it is possible to provide a vibrating feeder control apparatus with simple wiring and high reliability, which is capable of controlling a drive frequency at which a phase difference between a voltage and a displacement becomes a predetermined relationship even when no displacement sensor is provided on the vibrating feeder main body side, and an inexpensive vibrating feeder using this control apparatus. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a configuration diagram schematically showing a vibrating feeder according to an embodiment of the present invention. 
         FIG. 2  is an explanatory view schematically showing an electromagnetic drive unit provided in the vibrating feeder. 
         FIG. 3  is an explanatory view showing a relationship when voltage and current supplied to the electromagnetic drive unit are enlarged. 
         FIG. 4  is an explanatory diagram showing a relationship between voltage and current supplied to the electromagnetic driving unit and displacement for each vibration mode. 
         FIG. 5  is an explanatory view showing a case where voltage and current supplied to the electromagnetic driving unit are enlarged in the vicinity of a reference phase angle. 
         FIG. 6  is an explanatory view showing a way of thinking about correction of a drive frequency of the electromagnetic drive unit. 
         FIG. 7  is an explanatory diagram schematically showing an example of actual waveforms of voltage and current. 
         FIG. 8  is an explanatory diagram showing an example, when voltage and current are actually measured. 
     
    
    
     MODE FOR CARRYING OUT THE INVENTION 
     Hereinafter, one embodiment of the present invention will be described with reference to the drawings. 
     As shown in  FIG. 1 , a vibrating feeder Fv of this embodiment is configured as a so-called linear feeder comprising a vibrating feeder main body  1  and a vibrating feeder control apparatus  2  (hereinafter referred to as a “control apparatus  2 ”). 
     The vibrating feeder main body  1  can convey a work (not shown) placed on a movable body  12  by vibrating the movable body  12  in a longitudinal direction (width direction of the paper). That is, the longitudinal direction of the movable body  12  is set in the same direction as the work conveying direction. 
     The vibrating feeder main body  1  is configured as follows to vibrate the movable body  12 . 
     The vibrating feeder main body  1  includes a base  11  installed on a floor surface FL and a movable body  12  connected to the base  11  via a pair of leaf springs  13 ,  13  which are elastic support means. An elastic support means such as an anti-vibration rubber or the like may be provided between the base  11  and the floor surface FL. The leaf springs  13 ,  13  is disposed parallel to each other while being separated in the longitudinal direction of the movable body  12  (in the left-right direction of the paper) so that they are inclined slightly upward. Therefore, the movable body  12  is elastically supported on the base  11  while being displaceable in a direction perpendicular to the surfaces of the leaf springs  13 ,  13 , that is, in a slightly inclined direction including components in the longitudinal and vertical directions of the movable body  12 . 
     The vibrating feeder main body  1  further includes an electromagnetic drive unit De to be able to vibrate the movable body  12  in the above-described displaceable direction. Specifically, the, electromagnetic drive unit De comprises an electromagnet  14  and a magnetic core  15 . The electromagnet  14  is provided on the base  11  through a bracket  16  to be disposed with a magnetic attraction surface  14  an orthogonal to a horizontal direction. The magnetic core  15  has a rectangular plate shape and is fixed to the lower surface of the movable body  12  to extend downward. Being configured as above, the electromagnet  14  and the magnetic core  15  are disposed opposite to each other to generate a magnetic attraction force therebetween by a current flowing in the electromagnet  14 , thereby displacing the movable body  12 . 
       FIG. 2  is an enlarged view of the electromagnetic drive unit De.  FIG. 2( b )  shows a state viewed from the same direction as  FIG. 1 .  FIG. 2( a )  shows a state where the electromagnet  14  is viewed from the opposite side of the magnetic core  15 . 
     The electromagnet  14  comprises an iron core  14 A and a coil  14 B. In  FIG. 2( a ) , the coil  14 B is omitted, and in  FIG. 2( b ) , it is indicated by an imaginary line (two-dot chain line). The magnetic core  15  and the iron core  14 A are formed by laminating and integrating silicon steel plates that are ferromagnetic materials. 
     The iron core  14 A is formed to have an E-shape in a side view, and includes a rectangular back plate  16  extending in the vertical direction, a center protrusion  17  having a rectangular shape in a plan view extending to the magnetic core  15  from the center in the vertical direction, and a pair of outer protrusions  18 ,  18  having a rectangular shape in a plan view and extending toward the magnetic core  15  from the upper and lower ends of the back plate  16 . 
     Between the center protrusion  17  and each of the outer protrusions  18 ,  18 , two internal spaces Sp and Sp are formed opening sideways and toward the magnetic core  15 . The coil  14 B is configured to wind around the center protrusion  17  while straddling the two internal spaces Sp, Sp. 
     Being configured as above, when an electric current flows in a coil  14 B, as shown by an arrow in the figure, two magnetic paths M are formed to pass through the inside of the magnetic core  15  from the center projection part  17  and return to the center protrusion  17  via the outer protrusion  18  and the back plate  16 . Thereby a magnetic attraction force is generated between the electromagnet  14  and the magnetic core  15 . In order to form the magnetic path M in such a direction, it is necessary to make the current flowing through the coil  14 B counterclockwise as viewed from the magnetic core  15  side. When a current flows in the opposite direction, a magnetic path is formed in the opposite direction to the above. 
     Returning to  FIG. 1 , a sinusoidal AC voltage is applied from the control apparatus  2  to the electromagnet  14  constituting the electromagnetic drive part De, and a current corresponding to this flows, whereby a sinusoidal magnetic attraction force is generated between the base  11  and the movable body  12 , and this magnetic attraction force becomes an exciting force for the movable body  12  to cause a vibration in the movable body  12 . 
     The control apparatus  2  for controlling the vibrating feeder main body  1  configured as described above is configured as follows. 
     First, the control apparatus  2  includes an information processing unit  3 , an amplifier  4  that amplifies a PWM signal output from the information processing unit  3  to generate a drive voltage and supplies the drive voltage to the electromagnet  14 , and a current detector  5  that detects a current flowing from the amplifier  4  to the electromagnet  14 . 
     The information processing unit  3  is constituted by an ordinary microprocessor or the like having a CPU, a memory, and an interface. The memory has previously stored a program necessary for processing. The CPU successively takes out and executes a necessary program to achieve a desired function by cooperating with peripheral hardware resources. 
     The information processing unit  3  includes a storage unit  31 , a frequency setting unit  32 , a PWM signal generation unit  33 , a current detection unit  34 , a first current change rate generation unit  35 , a second current change rate generation unit  36 , and a frequency correction unit  37 , so that so-called resonance point follow-up control can be performed by cooperation of these components even thought a displacement detection signal is not input from the vibrating feeder main body  1  side. 
     The storage unit  31  stores an initial setting frequency f 0  for driving the electromagnetic drive unit De on startup, first and second reference phase angles θ 1  and θ 2  to be described later, a threshold value ΔRth of a current change rate difference ΔR used for frequency correction, a frequency correction amount Δf per one time, and a mask time Tm and the likes. 
     The frequency setting unit  32  sets a drive frequency f for driving the electromagnetic drive unit De, and outputs the drive frequency f to the PWM signal generation unit  33 . The frequency setting unit  32  reads an initial set frequency f 0  stored in the storage unit  31  and uses the value as the drive frequency f upon startup of operation, and sequentially updates the drive frequency f based on a frequency correction value input from the frequency correction unit  37  described later after entering steady operation. 
     Based on the drive frequency f input from the frequency setting unit  32 , the PWM signal generation unit  33  generates a PWM signal to obtain a sinusoidal pseudo AC voltage signal corresponding to the drive frequency f. The PWM signal consists of a rectangular positive voltage pulse signal and a negative voltage pulse signal as viewed to expand in minute time units. A pseudo AC voltage signal is generated by output of these pulse signals while changing a duty ratio, that is, while changing a pulse width. As described above, the PWM signal is amplified by the amplifier  4  and supplied to the electromagnet  14  as a drive voltage. 
     The current detection unit  34  can detect a current value flowing in the electromagnet  14  in real time by an input from the current detector  5  and output it as a current detection value. 
     The first current change rate generation unit  35  generates a first current change rate R θ1  at a first reference phase angle θ 1  from the current detection value detected by the current detection unit  34 . The second current change rate generation unit  36  generates a second current change rate R θ2  at a second reference phase angle θ 2  from the current detection value detected by the current detection unit  34 . 
     The frequency correction unit  37  compares absolute values |R θ1 |, |R θ2 | of the current change rates R θ1 , R θ2  generated by the current change rate generation units  35 ,  36 . Based on the comparison result, the frequency correction unit  37  determines whether or not to correct the drive frequency f and further whether to increase or decrease a current value of the drive frequency f when correcting the drive frequency f, and outputs a corresponding frequency correction value Δf or −Δf. 
     Specifically, when the current change rate difference ΔR obtained by subtracting the absolute value |R θ2 | of the second current change rate R θ2  from the absolute value |R θ1 | of the first current change rate R θ1  is determined to be within a range ±ΔR th  set by the predetermined threshold value ΔR th , since it is in the resonance state, the frequency correction unit  37  outputs a zero frequency correction value to the frequency setting section  32  without requiring frequency correction. In that case, the frequency setting unit  32  maintains a current value of the drive frequency f without making corrections. 
     When the current change rate difference ΔR is determined to be smaller than the above range, that is, smaller than −ΔR th , the frequency correction unit  37  outputs the frequency correction value Δf to the frequency setting unit  32  to add the frequency correction amount Δf for one time stored in the storage unit  31  to a current value of the drive frequency f as determined in advance. In that case, the frequency setting unit  32  corrects the drive frequency f to set a new drive frequency f+Δf. 
     Further, when the current change rate difference ΔR is determined to exceed the above range, that is, larger than ΔR th , the frequency correction unit  37  outputs the frequency correction value −Δf to the frequency setting unit  32  to subtract the frequency correction value Δf for one time from a current value of the drive frequency f. In that case, the frequency setting unit  32  corrects the drive frequency f to set a new drive frequency f−Δf. 
     Here, in order to explain the operation of the control apparatus  2  configured as described above, the principle of the resonance point follow-up control made by the control apparatus  2  will be described. 
     When a voltage V is applied to the electromagnetic drive unit De configured as shown in  FIG. 2 , a relation as shown by the following equation occurs between the current change rates di/dt (=R) representing a gradient of a current flowing in the coil  14 B. 
         di/dt=V/L    (Equation 1)
 
     Here, dt is a minute time, di is a current change value during a minute time dt, and L is an inductance. From Equation 2, it can be seen that the current change rate di/dt is inversely proportional to the inductance L when the voltage is constant. 
     Further, the inductance L has the following relation with the magnetic path M through which a magnetic flux passes by the electromagnet  14 . 
         L=μ   0   ·S·N   2 /( l   g   +l 1 c /μ r )   (Equation 2)
 
     Here, μ 0  so is a permeability in vacuum (=4π×10 −7 ), μ r  is a relative magnetic per (=15000) of a silicon steel plate forming the iron core  14 A and the magnetic core  15 , l c  is a length of the magnetic path M, S is a cross-sectional area of the iron core  14 A. 
     As shown in  FIG. 2 , it is assumed that the thickness of the magnetic core  15  is A, the length of the center protrusion  17  and the outer protrusion  18  is B, the thickness of the back plate  18  is C, the thickness of the outer protrusion  18  is D, and the thickness of the center protrusion  17  is set to 2×D. Further, it is assumed that the space between the center protrusion  17  and the outer protrusion  18  is E, and the width dimension of the core  14 A and the magnetic core  15  is F. In that case, if a gap formed between the iron core  14 A constituting the electromagnet  14  and the magnetic core  15  is l g , the length l c  of the magnetic path M can be obtained from the following equation. 
         l   c   =A+ 2 B+C+ 2 D+ 2 E+ 2 l   g    (Equation 3)
 
     Further, a sectional area S of the iron core  14 A can be obtained from the following equation. 
         S=D×F    (Equation 4)
 
     As can be seen from Equation 2, the inductance L is influenced by the permeability μ 0  of the vacuum, the sectional area S, the number of turns N, the gap l g , the magnetic path length l c , and the relative permeability μ r . Among them, a variable is only the gap lg and the magnetic path length lc. According to Equation 3, an influence of a change in the gap lg that can be given to the inductance L is much greater than an influence of a change in the magnetic path length lc. That is, the change in the inductance L occurs mostly due to the change in the gap lg. 
     Therefore, if a voltage is constant, the inductance L can be obtained from the current change rate di/dt according to Equation 1, and the gap l g  can be obtained by Equation 2. Further, qualitatively, it can be said that the gap l g  is constant when the current change rate di/dt is constant,the gap l g  is small when the current change rate di/dt is relatively small, and the gap l g  is large when the current change rate di/dt is relatively large. Since a change in the gap l g  means a displacement of the movable body  12  (see  FIG. 1 ), by obtaining the current change rate di/dt in a certain minute time, a displacement of the movable body  12  can be obtained without requiring a special displacement sensor. 
     Further, since the positive or negative of the current change rate di/dt changes only depending on the positive or negative of the applied voltage, it has little meaning in obtaining the displacement of the movable body  12 . Therefore, a magnitude relationship of the current change rate di/dt may be determined by |di/dt| with an absolute value. 
     Generally, a sinusoidal AC voltage is applied to the electromagnetic drive unit De (see  FIG. 1 ) used for the vibrating feeder, and a pseudo AC voltage by PWM control is often used for such an AC voltage. The present embodiment adopts this system. 
       FIG. 3  shows waveforms of a pseudo AC voltage used in the present embodiment and a current to flow in the electromagnet  14  when this voltage is applied. A part of the waveform is shown on the right side, and a magnified one is shown on the left side. 
     In the PWM control, a constant voltage of rectangular pulse shape is output by changing the pulse width and further by reversing positive and negative every half cycle. By collecting such a pulsed voltage, a sinusoidal pseudo AC voltage is created. Incidentally, the pulse width becomes the largest at a point of phase angle 90° where a positive peak value of a sine wave is obtained or at a point of phase angle of 270° where a negative peak value is obtained. On the contrary, the pulse width becomes the smallest at points of phase angles 0° and 180°. 
     By applying the pseudo AC voltage created in this way, a current flows only in one direction with respect to the electromagnet  14 , and the current value sinusoidally changes at the same frequency as a voltage. At this time, the current is accompanied by a phase angle delay of 90° with respect to the voltage. In order to obtain such a relationship between the voltage and the current, the known circuit configuration as described in Japanese Patent No. 4032192 can be effectively utilized. 
     When viewing the voltage and current having such a relationship by enlarging a minute time, the voltage forms a rectangular wave pulse voltage, and the current changes stepwise corresponding to the pulse voltage. For example, when paying attention to the point of phase angle 90° corresponding to the positive peak value of the voltage, the voltage is constant while one pulse voltage is being applied, and a gap l g  of the electromagnetic drive unit De hardly changes and can be regarded as constant in the minute time dt during which this pulse voltage is obtained. Thus, the current change rate R (=di/dt) during that time is substantially constant. Likewise, even when paying attention to the point of phase angle 270° corresponding to the negative peak value of the voltage, the current change rate R (=di/dt) during that time is substantially constant. 
     However, as will be described in detail later, when an AC voltage is applied to the electromagnetic drive unit De (see  FIG. 1 ) for vibration, the movable body  12  becomes a resonance state, and the phases of the AC voltage and displacement re reversed 180°. Therefore, the absolute value |R| (=|di/dt|) of the current change rate increases at the phase angle of 90°, because as the movable body  12  displaces in a negative direction and the gap l g  increase, and the absolute value |R| (=|di/dt|) of the current change rate decreases at the phase angle is 270°, because the movable body  12  displaces in a positive direction and the gap l g  decreases. 
     Here, the relationship between the voltage applied to the electromagnetic drive unit De and the displacement of the movable body  12  will be described.  FIG. 4  illustrates how the phase of displacement changes in accordance with a vibration form of the movable body  12 . 
     When a sinusoidal AC voltage as shown in  FIG. 4( a )  is applied to the electromagnetic drive unit De (see  FIG. 1 ), as shown in  FIG. 4( b ) , a current whose phase is delayed by 90° from the voltage flows. Since a magnetic attraction force proportional to a current is generated in the electromagnetic drive unit De, the movable body  12  is vibrated by this magnetic attraction force with the same phase as the current. Then, as shown in  FIG. 4( c ) , when a vibration form of the movable body  12  is in a forced vibration state, the displacement of the movable body  12  changes with the same phase as an exciting force, that is, the current. When the vibration form is in a damped vibration state as shown in  FIG. 4( e ) , the displacement of the movable body  12  changes with a phase of 180° opposite to the current. Furthermore, in a resonance state as shown in  FIG. 4( d ) , the displacement of the movable body  12  changes with a phase of 90° delayed from the current (exciting force). 
     Viewing this as a relationship between the voltage and displacement, the displacement phase delays by 90° with respect to the voltage during the forced vibration, the displacement phase delays by 270° with respect to the voltage during the damped vibration, and the displacement phase delays by 180° with respect to the voltage during the resonance. 
     Therefore, when setting a first reference phase angle θ 1  and a second reference phase θ 1  at positions symmetrical with respect to a phase angle θp at which a voltage peak value is obtained (hereinafter referred to as a “peak phase angle θp”), for example θp=90°, that is, when setting the first and second reference phase angles θ 1 , θ 2  to satisfy the relationship of θ 1 =θp−Δθ, θ 2 =θp+Δθ, a displacements at these positions are equal at resonance and different at forced vibration and damped vibration. 
     Considering this fact in combination with the above-described relation with the current change rate R (=di/dt), the following relation is obtained.  FIG. 5  shows the relationship between the voltage and the current at the first and second reference phase angles θ 1  and θ 2 . 
     As can be seen from the figure, since the first and second reference phase angles θ 1  and θ 2  are located symmetrically with respect to the peak phase angle θp, the pulse widths are substantially the same. Furthermore, since the momentary displacement values are equal at the first and second reference phase angles θ 1 , θ 2 , the current change rate R (=di/dt) corresponding to this pulse width is also equal. Therefore, the current change amount di corresponding to one pulse voltage is also equal. 
     The table shown in the upper part of  FIG. 6  shows a theoretical relationship between the first current change rate R θ1 , which is a current change rate R at the first reference phase angle θ 1 , and the second current change rate R θ2 , which is a current change rate R at the second reference phase angle θ 2 , in each case when the drive frequency f for driving the movable body  12  (see  FIG. 1 ) coincides with the resonance frequency, the case when it is lower than the resonance frequency, and the case when it exceeds the resonance frequency. The description shown in the lower part of  FIG. 6  shows a relationship to be used in actual control based on the above theoretical relationship. 
     Specifically, as shown in the table in the upper part of  FIG. 6 , when the drive frequency f coincides with the resonance frequency, as described above, the phase difference between the current and the displacement is 90°, and the absolute value |R θ1 | of the first current change rate is equal to the absolute value |R θ2 | of the second current change rate. When performing the resonance point follow-up control, correction of the drive frequency f is unnecessary in such a case. 
     When the drive frequency f deviates to less than the resonance frequency from this state, a displacement phase difference with respect to a current advances from 90°. At this time, the absolute value |R θ1 | of the first current change rate is smaller than the absolute value |R θ2 | of the second current change rate. Therefore, when performing the resonance point follow-up control, the drive frequency f may be corrected in an increasing direction to f+Δf. 
     On the other hand, when the drive frequency f deviates in a direction exceeding the resonance frequency, the displacement phase difference between a current and a displacement delays from 90°. At this time, the absolute value |R θ1 | of the first current change rate is larger than the absolute value |R θ2 | of the second current change rate. Therefore, when performing the resonance point follow-up control, the drive frequency f may be corrected in a decreasing direction to f−Δf. 
     In practice, however, it is difficult to shape a pulse voltage to a perfect rectangle, and it is difficult to change a current completely linearly while one pulse voltage is being output. 
       FIG. 7  schematically shows characteristics of the waveforms of actually obtained voltage and current. Like this, when switching a voltage signal from an off state (0) to an on state (+V), the voltage rises with a slight inclination due to a response delay immediately after the switching. Then, after maintaining a constant state, the voltage returns to the off state with a slight inclination. Likewise, a current rises dully due to a response delay immediately after the switching, and changes linearly with a certain inclination after gradually changing on a curve. Then, as the voltage switches to OFF, the current turns to decrease. 
     In other words, while a pulse signal is being output, the current change rate R (=di/dt) is not always constant. However, after a predetermined masking tame Tm has elapsed from immediately after the switching at which a pulse signal turns on, the inclination can be considered to be substantially constant, and the above-described relationship can be seen in this linear part. Therefore, a portion obtained by subtracting the mask time Tm from the pulse output time Tp is set as a minute time dt for generation of the current change rate R, and the current change rate di therebetween is obtained, and the current change rate R (=di/dt) may be generated from these values. 
       FIG. 8  shows enlarged waveforms of actually measured pulse voltage and current, and an example of a case where the current change rate R (=di/dt) is obtained by the above method from these waveforms. In these waveforms, since the first reference phase angle θ 1  and the second reference phase angle θ 2  are set within a range of ±90° around the phase angle 270° where the pseudo AC voltage is a negative peak value, the enlarged waveforms of the voltage and the current are upside down with respect to  FIG. 7 . 
     First,  FIG. 8( a )  shows enlarged waveforms of the voltage and current obtained when the drive frequency f is higher than the resonance frequency. At the first reference phase angle θ 1 , the current change amount is di=1.33 mA with respect to the minute time dt=22.8 μsec, and the absolute value of the first current change rate |R θ1 |=58.7 is obtained. At the second reference phase angle θ 2 , the current change amount is di=0.83 mA with respect to the minute time dt=21.7 μsec, an absolute value of the second current change rate |R θ2 |=38.3 is obtained. In this case, the current change rate difference ΔR=20.4 obtained by subtracting the absolute value |R θ2 | of the second current change rate from the absolute value |R θ1 | of the first current change rate is considerably large, about 35% of the absolute value |R θ1 | of the first current change rate. 
       FIG. 8( b )  shows enlarged waveforms of the voltage and current obtained when the drive frequency f coincides with the resonance frequency. At the first reference phase angle θ 1 , the current change amount is di=0.79 mA with respect to the minute time dt=21.3 μsec, and the absolute value |R θ1 |=37.3 of the first current change rate is obtained. At the second reference phase angle θ 2 , the current change amount is di=0.75 mA with respect to the minute time dt=22.0 μsec, an absolute value of the second current change rate |R θ2 |=34.1 is obtained. In this case, the current change rate difference ΔR=3.2 obtained by subtracting the absolute value |R θ2 | of the second current change rate from the absolute value |R θ1 | of the first current change rate is approximately 9% of the absolute value |R θ1 | of the first current change rate, that is almost zero. In other words, it can be said that the absolute values |R θ1 | and |R θ2 | of the current change rate are substantially equal. 
       FIG. 8( c )  shows enlarged waveforms of the voltage and current obtained when the drive frequency f is lower than the resonance frequency. At the first reference phase angle θ 1 , the current change amount is di=0.71 mA with respect to the minute time dt=19.2 μsec, and the absolute value of the first current change rate |R θ1 |=36.8 is obtained. At the second reference phase angle θ 2 , the current change amount is di=1.04 mA with respect to the minute time dt=20.7 μsec, and the absolute value of the second current change rate |R θ2 |=50.2 is obtained. In this case, the current change rate difference ΔR=−13.4 obtained by subtracting the absolute value |R θ2 | of the second current change rate from the absolute value |R θ1 | of the first current change rate is considerably small, about −36% of the absolute value |R θ1 | of the first current change rate. 
     According to the above actually measured waveforms, the mask time Tm is set to 10 to 15 μsec in the control apparatus  2  of the present embodiment. Further, in order to determine whether or not to correct the drive frequency f, the threshold value ΔRth is set for comparing the current change rate difference ΔR that is the difference between the absolute values |R θ1 |, |R θ2 | of each current change rate, and this threshold value ΔR th  is set to 10% of the absolute value |R θ1 | of the first current change rate. Moreover, when the current change rate difference ΔR is in a range of over −ΔR th  and below +ΔR th  that is set with zero interposed therebetween, the drive frequency f is not corrected, but corrected only when the current change rate difference is out of this range. More specifically, when the current change rate difference ΔR is less than −ΔR th , the drive frequency f is corrected to be high, and when the current change rate difference ΔR exceeds ΔR th , the drive frequency f is corrected to be low. 
     In the control apparatus  2 , even when no displacement sensor is provided in the vibrating feeder main body  1 , it is possible to operate while performing the resonance point follow-up control as described below using the above-described principle. 
     First, when driving of the vibrating feeder main body  1  is started by the control apparatus  2 , the frequency setting unit  32  reads the preset initial setting frequency f 0  set in the storage unit  31  and outputs it as a drive frequency f to the PWM signal generation unit  33 . The PWM signal generation unit  33  generates and outputs a PWM signal corresponding to the drive frequency f. The PWM signal is amplified by the amplifier  4  and supplied to the electromagnetic drive unit De as a pseudo AC voltage. As a result, the movable body  12  is vibrated by the drive frequency f to be able to convey the work placed on the movable body  12  by the vibration. As the initial setting frequency f 0 , it is preferable to use a resonance frequency of the vibrating feeder main body  1  in a non-loaded state or a final value of the drive frequency f at the previous driving. 
     Then, after shifting to a steady operation at the drive frequency f, the first current change rate generation unit  35  calculates the current variation amount di at the minute time dt, which is obtained by subtracting the preset mask time Tm from the output time Tp (see  FIG. 7 ,  FIG. 8 ) of the voltage signal for one pulse at the preset first reference phase angle θ 1 , by the current detection value from the current detection unit  34 , and generates the first current change rate R θ1  from these values. Similarly, the second current change rate generation unit  36  calculates the current change amount di at the minute time dt, which is obtained by subtracting the preset mask time Tm from the output time Tp of the voltage signal for one pulse at the preset second reference phase angle θ 2 , from the current detection value from the current detection unit  34 , and generates the second current change rate R θ2  from these values. 
     The first and second reference phase angles θ 1  and θ 2  are required to be set within a range of ±90° with respect to the phase angle θp at which the pseudo AC voltage has a peak value. Further, if it is set within a range of ±45°, the width of one pulse voltage constituting the PWM signal can be sufficiently obtained, and it is more preferable to obtain a highly accurate current change rate R. 
     On the basis of the first and second current change rates R θ1 , R θ2  obtained by the first and second current change rate generation units  35 ,  36  as described above, the frequency correction unit  37  determines whether or not to correct the drive frequency f, and decides a frequency correction value if a correction is to be performed. Specifically, in accordance with the concept described in the lower part of  FIG. 6 , the frequency correction unit  37  calculates the current change rate difference rate ΔR, which is obtained by subtracting the absolute value |Rθ 2 | of the second current change rate from the absolute value |R θ1 | of the first current change rate, and determines whether the obtained current change rate difference θR falls within the range of −ΔR th  or more and ΔR th  or less set by the threshold ΔR th  stored in the storage unit  31 . When the current change difference is within this range, an output that makes the frequency correction value zero is made to the frequency setting unit  32 , and the frequency setting unit  32  does not correct the drive frequency f. Further, when the current change rate difference ΔR is less than −ΔR th , the frequency correction amount Δf per one time stored in the storage unit  31  is outputted to the frequency setting unit  32  as a frequency correction value, and the frequency setting unit  32  makes a correction to update the drive frequency f to f+Δf. When the current change rate difference ΔR exceeds ΔR th , the frequency correction amount Δf per one time stored in the storage unit  31  is outputted to the frequency setting unit  32  as a frequency correction value, and the frequency setting unit  32  makes a correction to update the drive frequency f to f−Δf. 
     Although the threshold value ΔR th  is set to be about 10% of the first current change rate R θ1 , it may be smaller than this value. Further, the threshold value ΔRth may be obtained by calculation from the obtained first current change rate R θ1 . Further, the correction amount of the drive frequency f may be obtained by calculation to change in accordance with the magnitude of the current change rate difference ΔR. 
     Although the correction of the drive frequency f as described above is made for each cycle of the pseudo AC voltage, the correction may be made at an appropriate timing such as every 10 cycles. 
     By controlling the vibrating feeder main body  1  by using the control apparatus  2  in this way, even if the resonance frequency changes due to factors such as a weight change or unbalance of a work or a change with time of the characteristics of the leaf spring  13 , etc., it is possible to largely vibrate the movable body  12  with less energy by changing the drive frequency f to follow the resonance frequency change thereby properly conveying the work. 
     As described above, the vibrating feeder control apparatus  2  according to the present embodiment is used to drive the vibrating feeder main body  1  comprising the base  11 , the movable body  12  elastically supported by the base  11 , the electromagnet  14  provided on the base  11 , and the magnetic core  15  provided on the movable body  12  to oppose to the electromagnet  14 . The vibrating feeder control apparatus includes a PWM signal generation unit  33  that generates a PWM signal based on a set drive frequency f and applies a pseudo AC voltage corresponding to the PWM signal to the electromagnet  14 , a current detection unit  34  that detects a current flowing in the electromagnet  14  by the pseudo AC voltage, current change rate generation units  35 ,  36  that generate current change rates R θ1 , R θ2  at predetermined reference phase angles θ 1 , θ 2  within a cycle of the pseudo AC voltage based on the detection value by the current detection unit  34 , and a frequency correction unit  37  that corrects the drive frequency f based on the current change rates R θ1 , R θ2  at the reference phase angles θ 1 , θ 2  obtained by the current change rate generation units  35 ,  36 . 
     As being configured as described above, a pseudo AC voltage corresponding to the PWM signal generated by the PWM signal generation unit  33  is applied to the electromagnet  14 , and in a minute time, a pulse-like constant voltage is applied to the electromagnet  14 . When a constant voltage is applied to the electromagnet  14  as described above, the current change rate R (=di/dt) that is a gradient of a current flowing in the electromagnet  14  corresponds to the inductance L of the electromagnet  14 . Since the inductance L corresponds to the gap lg between the electromagnet  14  and the magnetic core  15 , in other words, the displacement amount of the movable body  12 , obtaining the current change rate R means the same as knowing the displacement amount of the movable body  12  at that time. Therefore, the current change rate generation units  35  and  36  generate the current change rates R θ1  and R θ2  at the predetermined reference phase angles θ 1  and θ 2 , and the frequency correction unit  37  corrects the drive frequency f based on the current change rates R θ1  and R θ2 , whereby it is possible to control the drive frequency f at which a phase difference between the voltage and the displacement becomes a predetermined relationship, ie, a phase difference of 180°, without using a displacement sensor for detecting the displacement of the movable body  12 . 
     Further, the first reference phase angle θ 1  and the second reference phase angle θ 2  are set as the reference phase angles θ 1 , θ 2  at substantially symmetrical positions around the peak phase angle θp that is a phase angle at which a peak value of the pseudo AC voltage is generated, the first current change rate generation unit  35  that generates the first current change rate R θ1  corresponding to the first reference phase angle θ 1  and the second current change rate generation unit  36  that generates the first current change rate R θ2  corresponding to the second reference phase angle θ 2  are provided as the current change rate generation units  35 ,  36 , and the frequency correction unit  37  is configured to correct the drive frequency f based on the first and second current change rates R θ1 , R θ2  obtained from the first and second current change rate generation units  35 ,  36 . Therefore, the drive frequency f can be corrected based on the two current change rates R θ1 , R θ2  obtained at positions substantially symmetrical around the peak phase angle θp so that the drive frequency f for driving the vibrating feeder main body  1  can be controlled with a higher accuracy. 
     Furthermore, the current change rate generation units  35 ,  36  are configured to generate the current change rates R θ1 , R θ2  during the period from turn on to turn off of the PWM signal for one pulse corresponding to the first and second reference phase angles θ 1  and θ 2 . Therefore, it is possible to accurately generate the current change rates R θ1  and R θ2  when a constant voltage is being applied and it is possible to make the control more easily. 
     Still further, the current change rate generation units  35 ,  36  generate the current change rates R θ1 , R θ2  during the period from turn on of the PWM signal for one pulse corresponding to the first and second reference phase angles θ 1  and θ 2  to turn off after the mask time Tm that is a predetermined time has elapsed. Therefore, it is possible to obtain the current change rates R θ1 , R θ2  more accurately by avoiding the influence of a response delay immediately after the switching. 
     Further, the first and second reference phase angles θ 1 , θ 2  are set within a range of ±90° around the peak phase angle θp at which a peak value of the pseudo AC voltage occurs, the frequency correction unit  37  is configured not to correct the drive frequency f when the current change rate difference ΔR obtained by subtracting the absolute value |R θ2 | of the second current change rate from the absolute value |R θ1 | of the first current change rate is in a predetermined range of −ΔR th  to R th , and correct the drive frequency f in a decreasing direction when the current change rate difference ΔR exceeds a predetermined range, that is, when it exceeds ΔR th , and correct the drive frequency f in an increasing direction when the current change rate difference ΔR is smaller than a predetermined range, that is, when it is lower than −ΔR th . Therefore, it is possible to easily determine whether the drive frequency f is substantially equal to the resonance frequency, or is large or small when the drive frequency f is deviated, and it is possible to easily correct the drive frequency f to be closer to the resonance frequency, and it is possible to suitably realize the resonance point follow-up control. 
     By configuring as a vibrating feeder Fv comprising the vibrating feeder control apparatus  2  and the vibrating feeder main body  1  controlled by the vibrating feeder control apparatus  2  as described above, it is possible to suitably perform the resonance point follow-up control without requiring a displacement sensor, and it is possible to realize an inexpensive vibrating feeder Fv with simple wiring and high reliability. 
     The specific structure of each part is not limited only to the embodiments described above. 
     Specifically, in the embodiment described above, the electromagnets  14  constituting the electromagnetic drive unit De is provided on the base  11  side and the magnetic core  15  is provided on the movable body  12  side, but contrary to this, the electromagnets  14  may be provided on the movable body  12  side and the magnetic core  15  may be provided on the base  11  side. 
     In the embodiment described above, a pseudo AC voltage is applied to flow a current only in one direction to the electromagnet  14 , but a current may flow while changing to positive and negative, even in this case, the effect similar to the above can be obtained. 
     Furthermore, in the embodiment described above, the drive frequency f is controlled to coincide with the resonance frequency. However, depending on the characteristics of the vibrating feeder main body  1 , the drive frequency f may be slightly shifted from the resonance frequency to obtain control stability. In that case, it is preferable that the first and reference phase angles θ 1 , θ 2  are not set to positions completely symmetrical with respect to the peak phase angle θp but are set to slightly shifted positions. By doing like this, it is possible to perform control by setting the phase difference between voltage and displacement to a predetermined relationship slightly shifted from 180°. 
     In the above embodiment, the vibrating feeder Fv is configured as a linear feeder, but it may be configured as a bowl feeder as disclosed in Patent Document 2. Similarly to the above, as long as the electromagnetic drive unit De is provided in the vibrating feeder main body  1 , the same effect can be obtained by using the control apparatus  2  with the same configuration as described above. 
     Various modifications are possible in other configurations without departing from the gist of the present invention. 
     DESCRIPTION OF REFERENCE NUMERALS 
       1  Vibrating feeder main body 
       2  Vibrating feeder control apparatus 
       11  Base 
       12  Movable body 
       14  Electromagnet 
       15  Magnetic core 
       33  PWM signal generation unit 
       34  Current detection unit 
       35  First current change rate generation unit 
       36  Second current change rate generation unit 
       37  Frequency correction unit 
     f Drive frequency 
     Fv Vibrating feeder 
     Δf Frequency correction amount per one time 
     L Inductance 
     R Current change rate (=di/dt) 
     R θ1  First current change rate 
     R θ2  Second current change rate 
     ΔR Current change rate difference 
     ΔR th  Threshold value of current change rate difference 
     Tm Mask time 
     θ 1  First reference phase angle 
     θ 2  Second reference phase angle 
     θp Peak phase angle