Patent Publication Number: US-8983408-B2

Title: Method and system for estimating and compensating non-linear distortion in a transmitter using calibration

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE 
     This application is a continuation of U.S. patent application Ser. No. 12/038,996 entitled “METHOD AND SYSTEM FOR ESTIMATING AND COMPENSATING FOR NON-LINEAR DISTORTION IN A TRANSMITTER USING CALIBRATION” and filed Feb. 28, 2008, the entirety of which is hereby incorporated by reference. 
    
    
     FIELD OF THE INVENTION 
     Certain embodiments of the invention relate to wireless communications. More specifically, certain embodiments of the invention relate to a method and system for estimating and compensating for non-linear distortion in a transmitter using calibration. 
     BACKGROUND OF THE INVENTION 
     A power amplification circuit in a wireless system is typically a large signal device. In wireless local area network (WLAN) systems, the power amplifier circuit may transmit output signals at average power levels in the range of 10 dBm to 21 dBm, and peak power levels of about 25 dBm, for example. In WLAN systems, which use orthogonal frequency-division multiplexing (OFDM) or complementary code keying (CCK) modulation, output power levels may vary widely such that the ratio of the peak power level to the average power level may be large, for example, 12 dB for OFDM and 6 dB for CCK. Because of these large swings in output power levels, power amplifier (PA) circuits may distort the output signal. Distortion, however, is a characteristic, which may be observed in PA circuits that are utilized across a wide range of applications, and may not be limited to PA circuits utilized in wireless systems. There are two metrics, which may be utilized to evaluate the distortion performance of PA circuits. These metrics may be referred to as amplitude modulation to amplitude modulation (AM-AM) distortion, and amplitude modulation to phase modulation (AM-PM) distortion. 
     The AM-AM distortion provides a measure of the output power level, p out , in response to the input power level, p in . In an ideal, non-distorting, PA circuit, the output power level changes linearly in response to a change in the input power level. The AM-AM distortion may be observed when, for example, the output power level does not change linearly in response to change in the input power level. 
     The AM-PM distortion provides a measure of the phase of the output signal in relation to the phase of the input signal (or phase change) in response to the input power level. The AM-PM distortion may be observed when, for example, the input to output phase-change varies in response to a change in input power level. 
     Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings. 
     BRIEF SUMMARY OF THE INVENTION 
     A method and system for estimating and compensating for non-linear distortion in a transmitter using calibration, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims. 
     These and other advantages, aspects and novel features of the present invention, as well as details of an illustrated embodiment thereof, will be more fully understood from the following description and drawings. 
    
    
     
       BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS 
         FIG. 1  is a block diagram illustrating an exemplary mobile terminal, which may be utilized in connection with an embodiment of the invention. 
         FIG. 2A  is a chart that illustrates exemplary AM-AM distortion in power amplifier circuitry, which may be utilized in connection with an embodiment of the invention. 
         FIG. 2B  is a chart that illustrates exemplary AM-PM distortion in power amplifier circuitry, which may be utilized in connection with an embodiment of the invention. 
         FIG. 3  is an exemplary block diagram illustrating a single chip wireless transmitter with compensation for nonlinear distortion, in accordance with an embodiment of the invention. 
         FIG. 4  is an exemplary block diagram illustrating a system for estimating and compensating for nonlinear distortion in a transmitter using calibration, in accordance with an embodiment of the invention. 
         FIG. 5  is a block diagram of an exemplary system for estimating and compensating for nonlinear distortion in a transmitter using calibration, in accordance with an embodiment of the invention. 
         FIG. 6  is a flowchart illustrating exemplary steps for estimating and compensating for nonlinear distortion in a transmitter, in accordance with an embodiment of the invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Certain embodiments of the invention may be found in a method and system for estimating and compensating for non-linear distortion in a transmitter using calibration. Various embodiments of the invention may comprise a single IC chip, which contains power amplifier circuitry (PA) in addition to circuitry for compensating for nonlinearity in the PA circuitry. Nonlinearity in the PA circuitry may be measured by AM-AM distortion and/or AM-PM distortion, for example. In various embodiments of the invention, AM-AM distortion may be determined by sending a range of input power levels, p in , to the PA circuitry and measuring the corresponding range of output power levels, p out , from the PA circuitry. The input power level may measure a radio frequency (RF) signal, which may be generated from a baseband signal. The baseband signal may comprise an In-Phase component, I BB , and a Quadrature-Phase component Q BB . In one aspect of the invention, AM-PM distortion may be measured by estimating a time delay from the input of the p in  signal to the PA circuitry to the output of the corresponding p out  signal. In another aspect of the invention, AM-PM distortion may be measured by estimating a relative change in the phase relationship between the corresponding baseband component signals I BB  and Q BB . 
       FIG. 1  is a block diagram illustrating an exemplary mobile terminal, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 1 , there is shown mobile terminal  120  that may comprise an RF receiver  123   a , an RF transmitter  123   b , a digital baseband processor  129 , a processor  125 , and a memory  127 . In some embodiments of the invention, the RF receiver  123   a , and RF transmitter  123   b  may be integrated into an RF transceiver  122 , for example. A single transmit and receive antenna  121  may be communicatively coupled to the RF receiver  123   a  and the RF transmitter  123   b . A switch  124 , or other device having switching capabilities may be coupled between the RF receiver  123   a  and RF transmitter  123   b , and may be utilized to switch the antenna  121  between transmit and receive functions. 
     The RF receiver  123   a  may comprise suitable logic, circuitry, and/or code that may enable processing of received RF signals. The RF receiver  123   a  may enable receiving RF signals in frequency bands utilized by various wireless communication systems, such as WLAN, Bluetooth, global system for mobile communications (GSM) and/or code division multiple access, for example. 
     The digital baseband processor  129  may comprise suitable logic, circuitry, and/or code that may enable processing and/or handling of baseband signals. In this regard, the digital baseband processor  129  may process or handle signals received from the RF receiver  123   a  and/or signals to be transferred to the RF transmitter  123   b  for transmission via a wireless communication medium. The digital baseband processor  129  may also provide control and/or feedback information to the RF receiver  123   a  and to the RF transmitter  123   b , based on information from the processed signals. The digital baseband processor  129  may communicate information and/or data from the processed signals to the processor  125  and/or to the memory  127 . Moreover, the digital baseband processor  129  may receive information from the processor  125  and/or the memory  127 , which may be processed and transferred to the RF transmitter  123   b  for transmission via the wireless communication medium. 
     The RF transmitter  123   b  may comprise suitable logic, circuitry, and/or code that may enable processing of RF signals for transmission. The RF transmitter  123   b  may enable transmission of RF signals in frequency bands utilized by various wireless communications systems, such as GSM and/or CDMA, for example. The RF transmitter  123   b  may contain PA circuitry, which may exhibit AM-AM and/or AM-PM distortion within its operating range. 
     The processor  125  may comprise suitable logic, circuitry, and/or code that may enable control and/or data processing operations for the mobile terminal  120 . The processor  125  may be utilized to control at least a portion of the RF receiver  123   a , the RF transmitter  123   b , the digital baseband processor  129 , and/or the memory  127 . In this regard, the processor  125  may generate at least one signal for controlling operations within the mobile terminal  120 . 
     The memory  127  may comprise suitable logic, circuitry, and/or code that may enable storage of data and/or other information utilized by the mobile terminal  120 . For example, the memory  127  may be utilized for storing processed data generated by the digital baseband processor  129  and/or the processor  125 . The memory  127  may also be utilized to store information, such as configuration information, which may be utilized to control the operation of at least one block in the mobile terminal  120 . For example, the memory  127  may comprise information necessary to configure the RF receiver  123   a  to enable receiving RF signals in the appropriate frequency band. 
       FIG. 2A  is a chart that illustrates exemplary AM-AM distortion in power amplifier circuitry, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 2A , the horizontal axis presents input power levels, P in , measured in units of decibels relative to 1 milliwatt (dBm). The input levels as shown in  FIG. 2A  range from −30 dBm to 30 dBm. The vertical axis presents Gain, measured in decibels (dB), which may be determined based on the ratio of output power levels, P out , to P in . 
     As shown in  FIG. 2A , the Gain level is approximately constant, at about 18 dB, over the range of levels P in  from about −30 dBm to about 5 dBm. The range of levels −30 dBm≦P in ≦5 dBm may correspond to the linear operating range for the PA circuitry. Over the range of levels P in &gt;5 dBm, the Gain level is shown to decrease with increasing levels P in . This range may correspond to a nonlinear operating range for the PA circuitry, and the decreasing Gain levels as a function of increasing levels P in , g(P in ), may be indicative of AM-AM distortion. 
       FIG. 2B  is a chart that illustrates exemplary AM-PM distortion in power amplifier circuitry, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 2B , the horizontal axis presents input power levels, P in , measured in units of decibels relative to 1 milliwatt (dBm). The input levels as shown in  FIG. 2A  range from −30 dBm to 30 dBm. The vertical axis presents Phase Shift, measured in degrees (°), which may be determined based on the phase difference between the P out  signal, measured at the output of the PA and the P in  signal, measured at the input of the PA. 
     As shown in  FIG. 2B , the Phase Shift level is approximately constant, at about 25°, over the range of levels P in  from about −30 dBm to about −10 dBm. Over the range of levels P in &gt;−10 dBm, the Phase Shift level is shown to vary with increasing levels P. The variations in Phase Shift with increasing levels P in , φ Δ (P in ), may be indicative of AM-PM distortion. 
       FIG. 3  is an exemplary block diagram illustrating a single chip wireless transmitter with compensation for nonlinear distortion, in accordance with an embodiment of the invention. Referring to  FIG. 3 , there is shown a single chip RF transmitter  300 . The single chip RF transmitter  300  may comprise a baseband processor  302 , a predistortion block  304 , a digital-to-analog converter (DAC) block  306 , an RF transmitter  308  and a power amplifier (PA)  312 . 
     The baseband processor  302  may comprise suitable logic, circuitry and/or code that may enable generation of baseband signals. The baseband signals may be represented in digital form and may comprise I BB  and Q BB  signal components. 
     The predistortion block  304  may comprise suitable logic, circuitry and/or code that may enable compensation for nonlinear distortion introduced by the PA  312 . The predistortion block  304  may enable compensation for nonlinear distortion by applying predistortion to an input baseband signal to generate a predistorted baseband signal. In this regard, the applied predistortion may comprise a gain predistortion function, g PD (P in ), and a phase shift predistortion function, φ PD (P in ). With reference to the exemplary Gain function shown in  FIG. 2A , the gain predistortion function may represented by the following relationship:
 
 g   PD ( P   in )=1  [1a]
 
for −30 dBm≦P in ≦5 dBm, and:
 
                       g   PD     ⁡     (     P   in     )       =       g   lin       g   ⁡     (         g   PD     ⁡     (     P   in     )       ·     P   in       )                 [     1   ⁢   b     ]               
for P in &gt;5 dBm, where g lin  represents the gain of the PA  312  across the linear operating region. The PA  312  gain, g(P in ), may be represented by the following relationship:
 
                     g   ⁡     (     P   in     )       =       P   out       P   in               [     1   ⁢   c     ]               
for an input power level to the PA  312 , P in , and an output power level from the PA  312 , P out .
 
     With reference to equation [1a], no gain predistortion may be applied to the input baseband signal when the PA  312  gain is substantially linear across a range of P in  levels. With reference to equation [1b], gain predistortion may be applied to the input baseband signal when the PA  312  gain is nonlinear across a range of P in  values. The gain predistortion may be in proportion to the nonlinearity of the PA  312  gain to enable compensation for AM-AM distortion in the PA  312 . Thus, for a given input baseband signal to the predistortion block  304 , s BB , a corresponding gain-predistorted baseband signal output from the predistortion block  304 , ŝ BB , may be represented by the following relationship:
 
 ŝ   BB   =g   PD ( P   in )· s   BB   [2]
 
     In various embodiments of the invention, the phase shift predistortion function, φ PD (P in ), may be additive to the Phase Shift function, φ Δ (P in ), such that the sum is a substantially constant value as shown in the following equation:
 
φ Δ ( P   in )+φ PD ( P   in )=φ const   ([3a]
 
where φ const  represents a reference phase shift value. Thus, a phase shift predistortion function, which may enable compensation for AM-PM distortion in the PA  312 , may be represented as follows:
 
φ PD ( P   in )=φ const −φ Δ ( P   in )  [3b]
 
     With reference to the exemplary Phase Shift function shown in  FIG. 2B , in an exemplary embodiment of the invention in which (I) φ const =45°, the phase predistortion applied to the input baseband signal may be represented by values:
 
φ PD ( P   in )&gt;0  [4a]
 
for P in &lt;0 dBm and values:
 
φ PD ( P   in )&lt;0  [4b]
 
for P in &gt;0 dBm.
 
     In various embodiments of the invention, phase shift predistortion may comprise altering the relative phase relationship between the input baseband signal components I BB  and Q BB . The amount of relative phase alteration may be determined based on equation [3b], for example. The relative phase shift alteration may comprise alteration of the amplitude ∥I BB ∥ relative to the amplitude ∥Q BB ∥, for example. 
     The DAC  306  may comprise suitable logic, circuitry, and/or code that may enable conversion of an input digital signal to a corresponding output analog signal representation. The RF transmitter  308  may comprise suitable logic, circuitry and/or code that may enable amplification of input signals to generate an amplified output signal. The amplified output signal may also be filtered, for example by a low pass filter (LPF). The RF transmitter  308  may be utilized in multistage amplifier systems wherein the output of the RF transmitter  308  may be an input to a subsequent amplification stage. 
     The PA  312  may comprise suitable logic, circuitry and/or code that may enable amplification of input signals, RF in , to generate output signals, RF out , of sufficient signal power (as measured by dBm, for example) for transmission via a wireless communication medium. The relationship between the input signal and the output signal may be represented as in the following equation:
 
RF out   =g ( P   in )·RF in   [5]
 
where P in  may represent the input power level for the input signal RF in . The PA  312  gain, g(P in ), may be constant with respect to P in  across a linear portion of the operating range of the PA  312 , and variable with respect to P in  across a nonlinear portion of the operating range. For example, with reference to  FIG. 2A , g(P in )=g lin  for the range of input power levels −30 dBm≦P in ≦5 dBm. For input power levels higher than 5 dBm, g(P in ) may decrease with increasing levels P in  as shown in  FIG. 2A .
 
     In an exemplary embodiment of the invention showing compensation for AM-AM distortion, the analog output signal, RF out , from the PA  312  may be substantially represented as a function of the digital input signal s BB  from the baseband processor  302  (after digital-to-analog conversion) as shown in the following equation: 
                     RF   out     ⁢     ←   DAC     ⁢       g   ⁡     (     P     i   ⁢   n       )       ·       g   PD     ⁡     (     P   in     )       ·     s   BB               [     6   ⁢   a     ]               
where the corresponding input signal to the PA  312 , RF in , may be represented by the following relationship:
 
                     RF   in     ⁢     ←   DAC     ⁢         g   PD     ⁡     (     P   in     )       ·     s   BB               [     6   ⁢   b     ]               
With reference to  FIG. 2A , for the input power level range −30 dBm≦P in ≦5 dBm (measured at the input of the PA  312 ):
 
                       RF   out     ⁢     ←   DAC     ⁢       g   lin     ·   1   ·     s   BB         ⁢     
     ⁢       RF   out     ⁢     ←   DAC     ⁢       g   lin     ·     s   BB                 [   7   ]               
for the input power level range P in &gt;5 dBm:
 
     
       
         
           
             
               
                 
                   
                     
                       RF 
                       out 
                     
                     ⁢ 
                     
                       ← 
                       DAC 
                     
                     ⁢ 
                     
                       
                         g 
                         ⁡ 
                         
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                             P 
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                           ) 
                         
                       
                       · 
                       
                         
                           g 
                           lin 
                         
                         
                           g 
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                             ( 
                             
                               P 
                               in 
                             
                             ) 
                           
                         
                       
                       · 
                       
                         s 
                         BB 
                       
                     
                   
                   ⁢ 
                   
                     
 
                   
                   ⁢ 
                   
                     
                       RF 
                       out 
                     
                     ⁢ 
                     
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                       DAC 
                     
                     ⁢ 
                     
                       
                         g 
                         lin 
                       
                       · 
                       
                         s 
                         BB 
                       
                     
                   
                 
               
               
                 
                   [ 
                   8 
                   ] 
                 
               
             
           
         
       
     
     In an exemplary embodiment of the invention showing compensation for AM-PM distortion, the phase of the analog output signal, RF out , from the PA  312  may be represented as φ(P out ) and the phase of the analog input signal, RF in , to the PA  312  may be represented as φ(P in ). The relative phase shift function φ Δ (P in ) may be represented by the following relationship:
 
φ Δ ( P   in )=φ( P   out )−φ( P   in )  [9]
 
The corresponding phase shift predistortion function, which may enable compensation for AM-PM distortion, may be represented by the following relationship:
 
φ PD ( P   in )=φ const −φ( P   out )+φ( P   in )  [10]
 
     Equations [4]-[10] present an exemplary method for compensating nonlinear distortion in a transmitter, in accordance with an embodiment of the invention. The nonlinear distortion compensation may comprise a gain predistortion function and/or a phase shift predistortion function. 
     In various embodiments of the invention, the predistortion function may comprise a gain predistortion function, g PD (P in ), and a phase shift predistortion function, e j·φ     PD     (P     in     ) , which may be applied to the I BB  and Q BB  components of the input baseband signal to generate predistorted signal components, I PD  and Q PD  as shown in the following equation:
 
 I   PD =Re{ g   PD ( P   in )· e   j·φ     PD     (P     in     ) ·( I   BB   +j·Q   BB )}
 
 Q   PD =Im{ g   PD ( P   in )· e   j·φ     PD     (P     in     ) ·( I   BB   +j·Q   BB )}  [11]
 
     Based on equation [11] a gain factor, g(P in ), may be determined, which may be computed based on the gain predistortion function and the phase shift predistortion function:
 
 g ( P   in )= g   PD ( P   in )· e   j·φ     PD     (P     in     )   [12]
 
       FIG. 4  is an exemplary block diagram illustrating a system for estimating and compensating for nonlinear distortion in a transmitter using calibration, in accordance with an embodiment of the invention. Referring to  FIG. 4 , there is shown a baseband samples block  402 , a power level computation block  404 , a dB conversion block  406  and a predistortion coefficient lookup table block  408 . 
     The baseband samples block  402  may comprise suitable logic, circuitry and/or code that may allow generation of a digital baseband signal. The digital baseband signal may comprise I component samples, I BB , and Q component samples, Q BB . 
     The power level computation block  404  may comprise suitable logic, circuitry and/or code that may enable computing an input power level, P in , based on received input baseband samples I BB  and Q BB . The power level may be computed by computing values ∥I BB ∥ 2  and ∥Q BB ∥ 2 , for example. In an exemplary embodiment of the invention, the computed power level may be expressed in units of milliwatts (mW). 
     The dB conversion block  406  may comprise suitable logic, circuitry and/or code that may enable conversion of an input power level, as expressed for example in units of mW, to units of dB or dBm. 
     The predistortion coefficient lookup table block  408  may comprise suitable logic, circuitry and/or code that may enable determining a predistortion coefficient value, ε, based on a received input power level expressed, for example, in dBm. The joint action of the gain predistortion function and phase shift predistortion function may enable modification of individual components, I BB  and Q BB , in the input baseband signal. 
     In operation, the baseband samples, I BB  and Q BB , output from the baseband samples block  402  may be multiplied the by predistortion coefficient, ε, to compute predistortion correction values, ΔI BB  and ΔQ BB , as shown in the following exemplary equations:
 
Δ I   BB =Rc{ε·( I   BB   +j·Q   BB )}
 
Δ Q   BB =Im{ε·( I   BB   +j·Q   BB )}  [13]
 
The predistortion correction values may then be added to the baseband samples to generate predistorted samples, I PD  and Q PD , as shown in the following exemplary equations:
 
 I   PD   =I   BB +ΔI BB  
 
 Q   PD   =Q   BB   +ΔQ   BB   [14]
 
The predistorted samples may be represented as in the following equation:
 
 I   PD =Re{ g ( P   in )·( I   BB   +Q   BB )}
 
 Q   PD =Im{ g ( P   in )·( I   BB   +Q   BB )}  [15]
 
In this regard, the gain factor, g(P in ), (as shown in equation [12]) may be represented as a function of the predistortion coefficient as shown in the following equation:
 
 g ( P   in )=1+ε  [16]
 
     In various embodiments of the invention, the predistorted samples, I PD  and Q PD , may comprise compensation for nonlinear distortion PA  312 , for example. The compensation may comprise adjustments in response to AM-AM distortion and/or AM-PM distortion. When the PA  312  is operating in the linear operating region, the predistortion coefficient values, ε, may be equal to 0, for example. In this case, the predistortion correction values, ΔI BB  and ΔQ BB , may also equal 0. Consequently, the predistorted samples, I PD  and Q PD , may be equal to the corresponding input baseband samples, I BB  and Q BB . 
     In various embodiments of the invention, the representation of the gain factor, g(P in ), as shown in equation [16] may enable avoidance of a loss of resolution in the computation of values for the gain factor when the values of ε are small in comparison to 1. 
       FIG. 5  is a block diagram of an exemplary system for estimating and compensating for nonlinear distortion in a transmitter using calibration, in accordance with an embodiment of the invention. Referring to  FIG. 5 , there is shown a transmitter  500 . The transmitter  500  may comprise a baseband samples block  502 , a scaler block  504 , a predistortion block  506 , a DAC block  507 , a PA block  508 , and analog-to-digital converter (ADC) block  510 , a scaler block  512  a summation block  514  and a coefficient update calculation block  516 . The DAC  507  is substantially similar to the DAC  306  described in  FIG. 3 . The PA  508  is substantially similar to the PA  312  also described in  FIG. 3 . 
     The baseband samples block  502  may comprise suitable logic, circuitry and/or code that may allow generation of a digital baseband signal. The digital baseband signal may comprise a plurality of samples, x. Each of the samples x may represent a complex number. 
     The scaler block  504  may comprise suitable logic, circuitry and/or code that may enable generation of scaled versions of received input samples x. The scaled samples output from the scaler block  504  may be represented as m·x, where m is a scale factor. In various embodiments of the invention, the value m may vary across a sequence of input samples. In various embodiments of the invention, the scaler block  504  may enable calibration by generating a sequence of scaled samples m·x, which comprise a range of signal amplitudes m·∥x∥. Each of the distinct scaled samples m·x may correspond to a distinct input power level P in . The scaled sample sequence may comprise an initial sequence portion, which comprises scaled samples m·x, which correspond to the linear operating range of the PA  508 . The maximum signal amplitude for which the PA  508  operates within the linear operating range may be referred to as ∥max_x lin ∥. Thus, the initial sequence portion may comprise scaled signal amplitudes m·x, which are less than or equal to ∥max_x lin ∥. The sequence of scaled samples, m·x, may comprise a subsequent sequence portion, which comprises signal amplitudes, which are greater than ∥max_x lin ∥. The subsequent sequence portion may comprise scaled samples m·x, which correspond to a nonlinear operating range of the PA  508 . 
     Across the range of signal amplitudes x in the sequence, the value of the scale factor may increase by a determined incremental value (or step size), γ, such that:
 
 m ( t+t   step )·ƒ x ( t+t   step )∥=(1+γ)· m ( t )(·∥ x ( t )∥  [17]
 
where t step  represents a time interval between increments in the value m based on the step size γ, x(t) represents a sample x generated at a time instant t (where each time unit may represent a time interval between generated samples), m(t) represents a scale factor value at the time instant t, m(t+t step ) represents the scale factor value at the time instant t+t step  and x(t+t step ) represents a sample generated at the time instant t+t step . The variable γ represents the iterative algorithm adaptation step size and is chosen to be small enough to enable algorithm convergence.
 
     The predistortion block  506  may comprise suitable logic, circuitry and/or code that may enable generation of predistorted versions of received input samples. When the predistortion block  506  receives as input a sequence of scaled samples m·x, the predistortion block  506  may output a corresponding sequence of predistorted samples m·c·x, where c is a predistortion scale factor. The value of the predistortion scale factor at a time instant t, c(t), may be determined based on the input power level, P in (t), which corresponds to the scaled input sample received at approximately the time instant t. The predistortion block  506  may comprise memory, which may enable storage of predistortion scale values. In an exemplary embodiment of the invention, the predistortion scale values may be stored in a lookup table (LUT). 
     In various embodiments of the invention, the determined input power level P in (t) may be utilized to determine a value for a predistortion coefficient, ε. The predistortion scale factor may then be determined as in the following equation:
 
 c= 1+ε  [18]
 
     In various embodiments of the invention, the predistortion block  506  may receive, as input, updated predistortion coefficient values corresponding to a current input power level P n (t). The updated predistortion coefficient value may replace the current predistortion coefficient value and may be utilized at a later time instant t′ in connection with a received input sample for which P in (t′)=P in (t). 
     The DAC  507  may receive digital predistorted input samples and generate an equivalent predistorted analog signal. The PA  508  may receive a predistorted analog input signal, RF in , and generate an amplified analog output signal, RF out  as shown in equation [5]. The value of the gain associated with the PA  508 , g, may reflect distortions due to AM-AM distortion and/or AM-PM distortion when the PA  508  is operating in a nonlinear operating range. 
     The ADC  510  may comprise suitable logic, circuitry, and/or code that may enable conversion of an input analog signal to a corresponding output digital signal representation. In various embodiments of the invention, the ADC  510  may receive an analog input signal, RF out , and generate a corresponding digital representation, m·c·g·x. 
     The scaler block  512  may be substantially similar to the scaler block  504 . In various embodiments of the invention, the scaler block  512  may receive input samples m·c·g·x, apply a scale factor k/m, and generate output samples k·c·g·x. In an exemplary embodiment of the invention, the value for the scale quantity k may be approximated to the known linear gain of complete transmit and receive path as follows: 
                   k   =     1     g   lin               [   19   ]               
where g lin  represents the gain factor for the PA  508  when the PA  508  is operating in a linear operating range.
 
     The output from the scaler block  512 , m·c·g·x, may be multiplied by the output of the baseband samples block  502 , x, to generate a correlation value k·c·g·x 2 . The correlation value may represent a correlation between the samples generated by the baseband samples block  502  and the digital representation of the scaled output signal RF out  generated by the PA  508 . 
     The summation block  514  may comprise suitable logic, circuitry and/or code that may enable summation of received input samples, k·c·g·x 2 , over a time duration, t sum . In various embodiments of the invention, the summation time duration t sum  may correspond to a portion of the time duration, t step , during which the value m(t) is constant. In this regard, the summation block  514  may compute an output, which represents a value, k·c·g·∥x∥ 2 , of the received input samples computed over the time duration t sum , where x represents values for samples x received over the time duration t sum . 
     In many transmitter systems, the output signal, RF out , generated by the PA  508  may contain noise, which was introduced during the process of generating the output signal. The level of noise may be estimated by computing a signal to noise ratio (SNR) for the output signal RF out . In various embodiments of the invention, the summation block  514  may enable the computation of a value k·c·g·∥x∥ 2  for which the SNR may be higher (or improved) relative to one or more of the instantaneous individual samples k·c·g·x 2  which may be received at the summation block  514  at various time instants across the time duration t sum . 
     The coefficient update calculation block  516  may comprise suitable logic, circuitry and/or code that may enable computation of updated coefficient values ε new  based on one or more received computed values k·c·g·∥x∥ 2 . In an exemplary embodiment of the invention, the updated coefficient value may be computed as shown in the following equation:
 
ε new =( Y   0   ·Y−Y   0   ·Y   ) *)·γ+ε current   [20]
 
where Y 0  represents the value, k·c·g·ƒx∥ 2 , which was computed at an amplitude value of the scaled baseband samples m·x for which the PA  508  is operating in the linear operating range. Y 0 * represents the complex conjugate of the complex quantity Y 0  and Y represents the value, k·c·g·∥x∥ 2 , which was computed across the current t sum  time duration. The variable γ represents the step size for increasing the scale factor m. The predistortion coefficient value ε current  represents the current value for the predistortion coefficient value, which may be utilized in the current t sum  time duration. The expression (Y 0 ·Y−Y 0 ·Y 0 *) represents a cross-correlation value between the correlation values Y and Y 0 . The computed value ε new  may be communicated to the predistortion block  506 , which may replace the current stored predistortion coefficient value, ε current , with the updated predistortion coefficient value ε new .
 
     In various embodiments of the invention, the transmitter  500  may comprise a calibration method and system, which may enable estimation of non linear distortion within the transmitter  500 . The method may also enable computation of predistortion scale factors, which may be utilized to enable compensation for nonlinear distortion within the transmitter. The calibration method may comprise generation of a sequence of samples, x, by the baseband samples block  502 . The samples may be scaled by the scaler  504  to generate a sequence of samples at varying amplitude levels and at varying input power levels P in . By varying the input power level based on a step size γ, the PA  508  may generate a range of output signals, RF out , when the PA  508  is operating within a linear operating range, and when the PA  508  is operating in a nonlinear operating range. By incrementing the scale value m in small steps γ, the digitized and scaled output from the PA  508  and the output from the baseband samples block  502  may be utilized to compute a current correlation measure k·c·g·x 2 . The current correlation measure may be computed across a time duration for which the value of the scale factor m is constant. The current computed correlation value and the previously computed correlation value may be utilized to determine an updated value ε new  for the predistortion coefficient ε current . The updated predistortion coefficient value may be stored as a replacement for ε current  in the predistortion block  506 . The predistortion block  506  may subsequently utilize the updated value when generating predistorted signals. 
     In various embodiments of the invention, initial values for the predistortion scale factors stored within the predistortion block  506  may each be equal to 1 for the full range of input power levels. During the calibration process, the stored initial values may be replaced with updated predistortion coefficient values ε new  across the range of input power levels. Upon completion of the calibration method, the predistortion block  506  may store a set of predistortion values ε, which may be utilized to compute predistortion scale factors, which may in turn enable compensation for nonlinearity in the PA  508  across the range of input power levels. 
     In various embodiments of the invention, the collocation of circuitry, which enables calibration, and circuitry, which enables compensation for nonlinear distortion in a single IC chip enable the predistortion estimates, determined during the calibration phase, to be subsequently utilized to compensate for nonlinear distortion in the transmitter  500  during normal operation. In an exemplary aspect of the invention, the coefficient update calculation block  516  may compute updated values for the predistortion coefficient ε new . The updated predistortion coefficient value may be communicated to the predistortion block  506 , wherein the updated predistortion coefficient value may be stored. The predistortion block  506  may subsequently utilize the updated predistortion coefficient to predistort subsequent baseband samples. 
       FIG. 6  is a flowchart illustrating exemplary steps for estimating and compensating for nonlinear distortion in a transmitter, in accordance with an embodiment of the invention. Referring to  FIG. 6 , steps  602 - 608  present steps for establishing a reference correlation value Y 0 , which may be computed when the PA  508  is operating in a linear operating range. In step  602 , the baseband samples block  502  may generate samples, x, for which the PA  508  is operating in the linear operating range. In step  604 , the processor  125  may configure the scaler  504  with a selected input scale factor, m, and the scaler  512  with a selected output scale factor k. In step  606 , the scaler  504  may scale the input baseband samples x. In step  608 , the summation block  514  may compute a correlation value Y 0 , which corresponds to the linear operating range of the PA  508 . 
     Steps  610 - 626  present steps for updating predistortion scale factors for input baseband samples when the PA  508  is operating in a nonlinear operating range. In step  610 , a step size γ may be selected by the processor  125 . In step  612 , an input scale factor corresponding to the current input power level, P in , may be selected by the processor. In step  614 , the baseband samples block  502  may generate baseband samples, x, for which the PA  508  may operate in a nonlinear operating range. In step  616 , the predistortion block  506  may predistort scaled baseband samples received from the scaler block  504 . The predistortion scale factor, c, may be determined based on the P in  value by reference to an LUT. 
     In step  618 , the output signal from the PA  508  may be utilized as a feedback signal. The current feedback signal may be converted from analog to digital format and scaled by the scaler  512 . In step  620 , the current scaled output value may be multiplied by the input baseband sample to generate a correlation value. A variable Y may be generated by summing a plurality of samples of the correlation value. In step  622 , the values Y, Y 0 , and γ may be utilized to compute an updated predistortion scale factor c new . The new predistortion scale factor may be utilized to update the LUT within the predistortion block  506  in step  624 . In step  626 , the value of the input scale factor m, may be incremented based on the value γ. Step  614  may follow step  626 . 
     Aspects of a system for estimating and compensating for non-linear distortion in a transmitter using calibration may include one more circuits that may enable estimation, within a single IC device, of distortion in output signals generated by a transmitter circuit  500 . The circuitry may enable compensation of the estimated distortion by predistorting subsequent input signals. The transmitter circuit  500  may generate subsequent output signals based on the predistorted subsequent input signals. The estimated distortion may comprise amplitude distortion (such as AM-AM distortion) and/or phase distortion (such as AM-PM distortion). A predistortion scale factor, c, may be computed that enables the predistortion of the subsequent input signals. The predistortion scale factor may be computed as shown in equation [18]. The predistortion coefficient value may be computed as shown in equation [20]. 
     Another embodiment of the invention may provide a machine-readable storage, having stored thereon, a computer program having at least one code section executable by a machine, thereby causing the machine to perform the steps as described herein for estimating and compensating for non-linear distortion in a transmitter using calibration. 
     Accordingly, the present invention may be realized in hardware, software, or a combination of hardware and software. The present invention may be realized in a centralized fashion in at least one computer system, or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system or other apparatus adapted for carrying out the methods described herein is suited. A typical combination of hardware and software may be a general-purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein. 
     The present invention may also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which when loaded in a computer system is able to carry out these methods. Computer program in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following: a) conversion to another language, code or notation; b) reproduction in a different material form. 
     While the present invention has been described with reference to certain embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted without departing from the scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the present invention without departing from its scope. Therefore, it is intended that the present invention not be limited to the particular embodiment disclosed, but that the present invention will include all embodiments falling within the scope of the appended claims.