Patent Publication Number: US-8996362-B2

Title: Device and method for a bandwidth extension of an audio signal

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a U.S. National Phase entry of PCT/EP2009/000329 filed Jan. 20, 2009, and claims priority to U.S. Patent Application No. 61/025,129 filed Jan. 31, 2008, and also claims period to German Patent Application No. 102008015702.3 filed Mar. 26, 2008, each of which is incorporated herein by references hereto. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates to the audio signal processing, and in particular, to the audio signal processing in situations in which the available data rate is rather small. 
     The hearing adapted encoding of audio signals for a data reduction for an efficient storage and transmission of these signals have gained acceptance in many fields. Encoding algorithms are known, in particular, as “MP3” or “MP4”. The coding used for this, in particular when achieving lowest bit rates, leads to the reduction of the audio quality which is often mainly caused by an encoder side limitation of the audio signal bandwidth to be transmitted. 
     It is known from WO 98 57436 to subject the audio signal to a band limiting in such a situation on the encoder side and to encode only a lower band of the audio signal by means of a high quality audio encoder. The upper band, however, is only very coarsely characterized, i.e. by a set of parameters which reproduces the spectral envelope of the upper band. On the decoder side, the upper band is then synthesized. For this purpose, a harmonic transposition is proposed, wherein the lower band of the decoded audio signal is supplied to a filterbank. Filterbank channels of the lower band are connected to filterbank channels of the upper band, or are “patched”, and each patched bandpass signal is subjected to an envelope adjustment. The synthesis filterbank belonging to a special analysis filterbank here receives bandpass signals of the audio signal in the lower band and envelope-adjusted bandpass signals of the lower band which were harmonically patched in the upper band. The output signal of the synthesis filterbank is an audio signal extended with regard to its bandwidth, which was transmitted from the encoder side to the decoder side with a very low data rate. In particular, filterbank calculations and patching in the filterbank domain may become a high computational effort. 
     Complexity-reduced methods for a bandwidth extension of band-limited audio signals instead use a copying function of low-frequency signal portions (LF) into the high frequency range (HF), in order to approximate information missing due to the band limitation. Such methods are described in M. Dietz, L. Liljeryd, K. Kjörling and O. Kunz, “Spectral Band Replication, a novel approach in audio coding,” in 112th AES Convention, Munich, May 2002; S. Meltzer, R. Böhm and F. Henn, “SBR enhanced audio codecs for digital broadcasting such as “Digital Radio Mondiale” (DRM),” 112th AES Convention, Munich, May 2002; T. Ziegler, A. Ehret, P. Ekstrand and M. Lutzky, “Enhancing mp3 with SBR: Features and Capabilities of the new mp3PRO Algorithm,” in 112th AES Convention, Munich, May 2002; International Standard ISO/IEC 14496-3:2001/FPDAM 1, “Bandwidth Extension,” ISO/IEC, 2002, or “Speech bandwidth extension method and apparatus”, Vasu Iyengar et al. U.S. Pat. No. 5,455,888. 
     In these methods no harmonic transposition is performed, but successive bandpass signals of the lower band are introduced into successive filterbank channels of the upper band. By this, a coarse approximation of the upper band of the audio signal is achieved. This coarse approximation of the signal is then in a further step approximated to the original by a post processing using control information gained from the original signal. Here, e.g. scale factors serve for adapting the spectral envelope, an inverse filtering and the addition of a noise carpet for adapting tonality and a supplementation by sinusoidal signal portions, as it is also described in the MPEG-4 Standard. 
     Apart from this, further methods exist such as the so-called “blind bandwidth extension”, described in E. Larsen, R. M. Aarts, and M. Danessis, “Efficient high-frequency bandwidth extension of music and speech”, In AES 112th Convention, Munich, Germany, May 2002 wherein no information on the original HF range is used. Further, also the method of the so-called “Artificial bandwidth extension”, exists which is described in K. Käyhkö, A Robust Wideband Enhancement for Narrowband Speech Signal; Research Report, Helsinki University of Technology, Laboratory of Acoustics and Audio signal Processing, 2001. 
     In J. Makinen et al.: AMR-WB+: a new audio coding standard for 3rd generation mobile audio services Broadcasts, IEEE, ICASSP &#39;05, a method for bandwidth extension is described, wherein the copying operation of the bandwidth extension with an up-copying of successive bandpass signals according to SBR technology is replaced by mirroring, for example, by upsampling. 
     Further technologies for bandwidth extension are described in the following documents. R. M. Aarts, E. Larsen, and O. Ouweltjes, “A unified approach to low- and high frequency bandwidth extension”, AES 115th Convention, New York, USA, October 2003; E. Larsen and R. M. Aarts, “Audio Bandwidth Extension—Application to psychoacoustics, Signal Processing and Loudspeaker Design”, John Wiley &amp; Sons, Ltd., 2004; E. Larsen, R. M. Aarts, and M. Danessis, “Efficient high-frequency bandwidth extension of music and speech”, AES 112th Convention, Munich, May 2002; J. Makhoul, “Spectral Analysis of Speech by Linear Prediction”, IEEE Transactions on Audio and Electroacoustics, AU-21(3), June 1973; U.S. patent application Ser. No. 08/951,029; U.S. Pat. No. 6,895,375. 
     Known methods of harmonic bandwidth extension show a high complexity. On the other hand, methods of complexity-reduced bandwidth extension show quality losses. In particular with a low bitrate and in combination with a low bandwidth of the LF range, artifacts such as roughness and a timber perceived to be unpleasant may occur. A reason for this is the fact that the approximated HF portion is based on a copying operation which leaves harmonic relations of the tonal signal portions unnoticed with regard to each other. This applies both, to the harmonic relation between LF and HF, and also to the harmonic relation within the HF portion itself. With SBR, for example, at the boundary between LF range and the generated HF range, occasionally rough sound impressions occur, as tonal portions copied from the LF range into the HF range, as for example illustrated in  FIG. 4   a , may now in the overall signal encounter tonal portions of the LF range as to be spectrally densely adjacent. Thus, in  FIG. 4   a , an original signal with peaks at  401 ,  402 ,  403 , and  404  is illustrated, while a test signal is illustrated with peaks at  405 ,  406 ,  407 , and  408 . By copying tonal portions from the LF range into the HF range, wherein in  FIG. 4   a  the boundary was at 4250 Hz, the distance of the two left peaks in the test signal is less than the base frequency underlying the harmonic raster, which leads to a perception of roughness. 
     As the width of tone-compensated frequency groups increases with an increase of the center frequency, as it is described in Zwicker, E. and H. Fastl (1999), Psychoacoustics: Facts and models. Berlin—Springerverlag, sinusoidal portions lying in the LF range in different frequency groups, by copying into the HF range, may come to lie in the same frequency group here, which also leads to a rough hearing impression as it may be seen in  FIG. 4   b . Here it is in particular shown that copying the LF range into the HF range leads to a denser tonal structure in the test signal as compared to the original. The original signal is distributed relatively uniformly across the spectrum in the higher frequency range, as it is in particular shown at  410 . In contrast, in particular in this higher range, the test signal  411  is distributed relatively non-uniformly across the spectrum and thus clearly more tonal than the original signal  410 . 
     SUMMARY 
     According to an embodiment, a device for a bandwidth extension of an audio signal may have: a signal spreader for generating a version of the audio signal as a time signal spread in time by a spread factor &gt;1; a decimator for decimating the temporally spread version of the audio signal by a decimation factor matched to the spread factor; a filter for extracting a distorted signal from the decimated audio signal containing a frequency range which is not contained in the audio signal, or for extracting a signal from the audio signal before a spreading by the signal spreader, wherein the signal contains a frequency range which is not contained in the audio signal after a spreading and decimation, wherein the distorted signal is distorted so that the distorted signal, the decimated audio signal, or the combination signal has a predetermined envelope; and a combiner for combining the distorted or undistorted signal with the audio signal to obtain an audio signal extended in its bandwidth. 
     According to another embodiment, a method for a bandwidth extension of an audio signal may have the steps of: generating a version of the audio signal as a time signal temporally spread by a spread factor &gt;1; decimating the temporally spread version of the audio signal by the decimation factor which is matched to the spread factor; extracting a distorted signal from the decimated audio signal containing a frequency range which is not contained in the audio signal, or extracting a signal from the audio signal before spreading, the signal containing a frequency range not contained in the audio signal after a spreading and decimation, wherein the distorted signal is distorted so that the extracted signal, the decimated audio signal or the combination signal has a predetermined envelope, and combining the distorted or undistorted signal with the audio signal to obtain an audio signal extended in its bandwidth. 
     Another embodiment may have a computer program having a program code for performing the above method for a bandwidth extension of an audio signal, when the computer program is executed on a computer. 
     The inventive concept for a bandwidth extension is based on a temporal signal spreading for generating a version of the audio signal as a time signal which is spread by a spread factor &gt;1 and a subsequent decimation of the time signal to obtain a transposed signal, which may then for example be filtered by a simple bandpass filter to extract a high-frequency signal portion which may only still be distorted or changed with regard to its amplitude, respectively, to obtain a good approximation for the original high-frequency portion. The bandpass filtering may alternatively take place before the signal spreading is performed, so that only the desired frequency range is present after spreading in the spread signal, so that a bandpass filtering after spreading may be omitted. 
     With the harmonic bandwidth extension on the one hand, problems resulting from a copying or mirroring operation, or both, may be prevented based on a harmonic continuation and spreading of the spectrum using the signal spreader for spreading the time signal. On the other hand, a temporal spreading and subsequent decimation may be executed easier by simple processors than a complete analysis/synthesis filterbank, as it is for example used with the harmonic transposition, wherein additionally decisions have to be made on how patching within the filterbank domain should take place. 
     For signal spreading, a phase vocoder may be used for which there are implementations of minor effort. In order to obtain bandwidth extensions with factors &gt;2, also several phase-vocoders may be used in parallel, which is advantageous, in particular with regard to the delay of the bandwidth extension which has to be low in real time applications. Alternatively, other methods for signal spreading are available, such as for example the PSOLA method (Pitch Synchronous Overlap Add). 
     In an embodiment of the present invention, the LF audio signal is first extended in the direction of time with the maximum frequency LFmax with the help of the phase vocoder, i.e. to an integer multiple of the conventional duration of the signal. Hereupon, in a downstream decimator, a decimation of the signal by the factor of the temporal extension takes place which in total leads to a spreading of the spectrum. This corresponds to a transposition of the audio signal. Finally, the resulting signal is bandpass filtered to the range (extension factor−1)·LFmax to extension factor·LFmax. Alternatively, the individual high frequency signals generated by spreading and decimation may be subjected to a bandpass filtering such that in the end they additively overlay across the complete high frequency range (i.e. from LFmax to k*LFmax). This is sensible for the case that still a higher spectral density of harmonics is desired. 
     The method of harmonic bandwidth extension is executed in an embodiment of the present invention in parallel for several different extension factors. As an alternative to the parallel processing, also a single phase vocoder may be used which is operated serially and wherein intermediate results are buffered. Thus, any bandwidth extension cut-off frequencies may be achieved. The extension of the signal may alternatively also be executed directly in the frequency direction, i.e. in particular by a dual operation corresponding to the functional principle of the phase vocoder. 
     Advantageously, in embodiments of the invention, no analysis of the signal is necessitated with regard to harmonicity or fundamental frequency. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       In the following, embodiments of the present invention are explained in more detail with reference to the accompanying drawings, in which: 
         FIG. 1  shows a block diagram of the inventive concept for a bandwidth extension of an audio signal; 
         FIG. 2   a  shows a block diagram of a device for a bandwidth extension of an audio signal according to an aspect of the present invention; 
         FIG. 2   b  shows an improvement of the concept of  FIG. 2   a  with transient detectors; 
         FIG. 3  shows a schematical illustration of the signal processing using spectrums at certain points in time of an inventive bandwidth extension; 
         FIG. 4   a  shows a comparison between an original signal and a test signal providing a rough sound impression; 
         FIG. 4   b  shows a comparison of an original signal to a test signal also leading to a rough auditory impression; 
         FIG. 5   a  shows a schematical illustration of the filterbank implementation of a phase vocoder; 
         FIG. 5   b  shows a detailed illustration of a filter of  FIG. 5   a;    
         FIG. 5   c  shows a schematical illustration for the manipulation of the magnitude signal and the frequency signal in a filter channel of  FIG. 5   a;    
         FIG. 6  shows a schematical illustration of the transformation implementation of a phase vocoder; 
         FIG. 7   a  shows a schematical illustration of the encoder side in the context of the bandwidth extension; and 
         FIG. 7   b  shows a schematical illustration of the decoder side in the context of a bandwidth extension of an audio signal. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  shows a schematical illustration of a device or a method, respectively, for a bandwidth extension of an audio signal. Only exemplarily,  FIG. 1  is described as a device, although  FIG. 1  may simultaneously also be regarded as the flowchart of a method for a bandwidth extension. Here, the audio signal is fed into the device at an input  100 . The audio signal is supplied to a signal spreader  102  which is implemented to generate a version of the audio signal as a time signal spread in time by a spread factor greater than 1. The spread factor in the embodiment illustrated in  FIG. 1  is supplied via a spread factor input  104 . The spread audio time signal present at an output  103  of the signal spreader  102  is supplied to a decimator  105  which is implemented to decimate the temporally spread audio time signal  103  by a decimation factor matched to the spread factor  104 . This is schematically illustrated by the spread factor input  104  in  FIG. 1 , which is plotted in dashed lines and leads into the decimator  105 . In one embodiment, the spread factor in the signal spreader is equal to the inverse of the decimation factor. If, for example, a spread factor of 2.0 is applied in the signal spreader  102 , a decimation with a decimation factor of 0.5 is executed. If, however, the decimation is described to the effect that a decimation by a factor of 2 is performed, i.e. that every second sample value is eliminated, then in this illustration, the decimation factor is identical to the spread factor. Alternative ratios between spread factor and decimation factor, for example integer ratios or rational ratios, may also be used depending on the implementation. The maximum harmonic bandwidth extension is achieved, however, when the spread factor is equal to the decimation factor, or to the inverse of the decimation factor, respectively. 
     In an embodiment of the present invention, the decimator  105  is implemented to, for example, eliminate every second sample (with a spread factor equal to 2) so that a decimated audio signal results which has the same temporal length as the original audio signal  100 . Other decimation algorithms, for example, forming weighted average values or considering the tendencies from the past or the future, respectively, may also be used, although, however, a simple decimation may be implemented with very little effort by the elimination of samples. The decimated time signal  106  generated by the decimator  105  is supplied to a filter  107 , wherein the filter  107  is implemented to extract a bandpass signal from the decimated audio signal  106 , which contains frequency ranges which are not contained in the audio signal  100  at the input of the device. In the implementation, the filter  107  may be implemented as a digital bandpass filter, e.g. as an FIR or IIR filter, or also as an analog bandpass filter, although a digital implementation may be of advantage. Further, the filter  107  is implemented such that it extracts the upper spectral range generated by the operations  102  and  105  wherein, however, the bottom spectral range, which is anyway covered by the audio signal  100 , is suppressed as much as possible. In the implementation, the filter  107  may also be implemented such, however, that it also extracts signal portions with frequencies as a bandpass signal contained in the original signal  100 , wherein the extracted bandpass signal contains at least one frequency band which was not contained in the original audio signal  100 . 
     The bandpass signal  108 , output by the filter  107 , is supplied to a distorter  109 , which is implemented to distort the bandpass signals so that the bandpass signal comprises a predetermined envelope. This envelope information which may be used for distorting may be input externally, and even come from an encoder or may also be generated internally, for example, by a blind extrapolation from the audio signal  100 , or based on tables stored on the decoder side indexed with an envelope of an audio signal  100 . The distorted bandpass signal  110  output by the distorter  109  is finally supplied to a combiner  111  which is implemented to combine the distorted bandpass signal  110  to the original audio signal  100  which was also distorted depending on the implementation (the delay stage is not indicated in  FIG. 1 ), to generate an audio signal extended with regard to its bandwidth at an output  112 . 
     In an alternative implementation, the sequence of distorter  109  and combiner  111  is inverse to the illustration indicated in  FIG. 1 . Here, the filter output signal, i.e. the bandpass signal  108 , is directly combined with the audio signal  100 , and the distortion of the upper band of the combined signal which is output from the combiner  111  is only executed after combining by the distorter  109 . In this implementation, the distorter operates as a distorter for distorting the combination signal so that the combination signal comprises a predetermined envelope. The combiner is in this embodiment thus implemented such that it combines the bandpass signal  108  with the audio signal  100  to obtain an audio signal which is extended regarding its bandwidth. In this embodiment, in which the distortion only takes place after combination, it is of advantage to implement the distorter  109  such that it does not influence the audio signal  100  or the bandwidth of the combination signal, respectively, provided by the audio signal  100 , as the lower band of the audio signal was encoded by a high-quality encoder and is, on the decoder side, in the synthesis of the upper band, so to speak the measure of all things and should not be interfered with by the bandwidth extension. 
     Before detailed embodiments of the present invention are illustrated a bandwidth extension scenario is illustrated with reference to  FIGS. 7   a  and  7   b , in which the present invention may be implemented advantageously. An audio signal is fed into a lowpass/highpass combination at an input  700 . The lowpass/highpass combination on the one hand includes a lowpass (LP), to generate a lowpass filtered version of the audio signal  700 , illustrated at  703  in  FIG. 7   a . This lowpass filtered audio signal is encoded with an audio encoder  704 . The audio encoder is, for example, an MP3 encoder (MPEG1 Layer 3) or an AAC encoder, also known as an MP4 encoder and described in the MPEG4 Standard. Alternative audio encoders providing a transparent or advantageously psychoacoustically transparent representation of the band-limited audio signal  703  may be used in the encoder  704  to generate a completely encoded or psychoacoustically encoded and psychoacoustically transparently encoded audio signal  705 , respectively. The upper band of the audio signal is output at an output  706  by the highpass portion of the filter  702 , designated by “HP”. The highpass portion of the audio signal, i.e. the upper band or HF band, also designated as the HF portion, is supplied to a parameter calculator  707  which is implemented to calculate the different parameters. These parameters are, for example, the spectral envelope of the upper band  706  in a relatively coarse resolution, for example, by representation of a scale factor for each psychoacoustic frequency group or for each Bark band on the Bark scale, respectively. A further parameter which may be calculated by the parameter calculator  707  is the noise carpet in the upper band, whose energy per band may be related to the energy of the envelope in this band. Further parameters which may be calculated by the parameter calculator  707  include a tonality measure for each partial band of the upper band which indicates how the spectral energy is distributed in a band, i.e. whether the spectral energy in the band is distributed relatively uniformly, wherein then a non-tonal signal exists in this band, or whether the energy in this band is relatively strongly concentrated at a certain location in the band, wherein then rather a tonal signal exists for this band. Further parameters consist in explicitly encoding peaks relatively strongly protruding in the upper band with regard to their height and their frequency, as the bandwidth extension concept, in the reconstruction without such an explicit encoding of prominent sinusoidal portions in the upper band, will only recover the same very rudimentarily, or not at all. 
     In any case, the parameter calculator  707  is implemented to generate only parameters  708  for the upper band which may be subjected to similar entropy reduction steps as they may also be performed in the audio encoder  704  for quantized spectral values, such as for example differential encoding, prediction or Huffman encoding, etc. The parameter representation  708  and the audio signal  705  are then supplied to a datastream formatter  709  which is implemented to provide an output side datastream  710  which will typically be a bitstream according to a certain format as it is for example normalized in the MPEG4 Standard. 
     The decoder side, as it is especially suitable for the present invention, is in the following illustrated with regard to  FIG. 7   b . The datastream  710  enters a datastream interpreter  711  which is implemented to separate the parameter portion  708  from the audio signal portion  705 . The parameter portion  708  is decoded by a parameter decoder  712  to obtain decoded parameters  713 . In parallel to this, the audio signal portion  705  is decoded by an audio decoder  714  to obtain the audio signal which was illustrated at  100  in  FIG. 1 . 
     Depending on the implementation, the audio signal  100  may be output via a first output  715 . At the output  715 , an audio signal with a small bandwidth and thus also a low quality may then be obtained. For a quality improvement, however, the inventive bandwidth extension  720  is performed, which is for example implemented as it is illustrated in  FIG. 1  to obtain the audio signal  112  on the output side with an extended or high bandwidth, respectively, and a high quality. 
     In the following, with reference to  FIG. 2   a , an implementation of the bandwidth extension implementation of  FIG. 1  is illustrated, which may be used in block  712  of  FIG. 7   b .  FIG. 2   a  firstly includes a block designated by “audio signal and parameter”, which may correspond to block  711 ,  712 , and  714  of  FIG. 7   b , and is designated by  200 . Block  200  provides the output signal  100  as well as decoded parameters  713  on the output side which may be used for different distortions, like for example for a tonality correction  109   a  and an envelope adjustment  109   b . The signal generated or corrected, respectively, by the tonality correction  109   a  and the envelope adjustment  109   b , is supplied to the combiner  111  to obtain the audio signal on the output side with an extended bandwidth  112 . 
     The signal spreader  102  of  FIG. 1  may be implemented by a phase vocoder  202   a . The decimator  105  of  FIG. 1  may be implemented by a simple sample rate converter  205   a . The filter  107  for the extraction of a bandpassed signal may be implemented by a simple bandpass filter  107   a . In particular, the phase vocoder  202   a  and the sample rate decimator  205   a  are operated with a spread factor=2. 
     A further “train” consisting of the phase vocoder  202   b , decimator  205   b  and bandpass filter  207   b  may be provided to extract a further bandpass signal at the output of the filter  207   b , comprising a frequency range between the upper cut-off frequency of the bandpass filter  207   a  and three times the maximum frequency of the audio signal  100 . 
     In addition to this, a k-phase vocoder  202   c  is provided achieving a spreading of the audio signal by the factor k, wherein k is an integer number greater than 1. A decimator  205  is connected downstream to the phase vocoder  202   c , which decimates by the factor k. Finally, the decimated signal is supplied to a bandpass filter  207   c  which is implemented to have a lower cut-off frequency which is equal to the upper cut-off frequency of the adjacent branch and which has an upper cut-off frequency which corresponds to the k-fold of the maximum frequency of the audio signal  100 . All bandpass signals are combined by a combiner  209 , wherein the combiner  209  may for example be implemented as an adder. Alternatively, the combiner  209  may also be implemented as a weighted adder which, depending on the implementation, attenuates higher bands more strongly than lower bands, independent of the downstream distortion by the elements  109   a ,  109   b . In addition to this, the system illustrated in  FIG. 2   a  includes a delay stage  211  which guarantees that a synchronized combination takes place in the combiner  111  which may for example be a sample-wise addition. 
       FIG. 3  shows a schematical illustration of different spectrums which may occur in the processing illustrated in  FIG. 1  or  FIG. 2   a . The partial image ( 1 ) of  FIG. 3  shows a band-limited audio signal as it is for example present at  100  in  FIG. 1 , or  703  in  FIG. 7   a . This signal may be spread by the signal spreader  102  to an integer multiple of the original duration of the signal and subsequently decimated by the integer factor, which leads to an overall spreading of the spectrum as it is illustrated in the partial image ( 2 ) of  FIG. 3 . The HF portion is illustrated in  FIG. 3 , as it is extracted by a bandpass filter comprising a passband  300 . In the third partial image ( 3 ),  FIG. 3  shows the variants in which the bandpass signal is already combined with the original audio signal  100  before the distortion of the bandpass signal. Thus, a combination spectrum with an undistorted bandpass signal results, wherein then, as indicated in the partial image ( 4 ), a distortion of the upper band, but if possible, no modification of the lower band takes place to obtain the audio signal  112  with an extended bandwidth. 
     The LF signal in the partial image ( 1 ) has the maximum frequency LFmax. The phase vocoder  202   a  performs a transposition of the audio signal such that the maximum frequency of the transposed audio signal is 2LFmax. Now, the resulting signal in the partial image ( 2 ) is bandpass filtered to the range LFmax to 2LFmax. Generally seen, when the spread factor is designated by k (k&gt;1), the bandpass filter comprises a passband of (k−1)·LFmax to k·LFmax). The procedure illustrated in  FIG. 3  is repeated for different spread factors, until the desired highest frequency k·LFmax is achieved, wherein k=the maximum extension factor kmax. 
     In the following, with reference to  FIGS. 5 and 6 , implementations for a phase vocoder  202   a ,  202   b ,  202   c  are illustrated according to the present invention.  FIG. 5   a  shows a filterbank implementation of a phase vocoder, wherein an audio signal is fed in at an input  500  and obtained at an output  510 . In particular, each channel of the schematic filterbank illustrated in  FIG. 5   a  includes a bandpass filter  501  and a downstream oscillator  502 . Output signals of all oscillators from every channel are combined by a combiner, which is for example implemented as an adder and indicated at  503 , in order to obtain the output signal. Each filter  501  is implemented such that it provides an amplitude signal on the one hand and a frequency signal on the other hand. The amplitude signal and the frequency signal are time signals illustrating a development of the amplitude in a filter  501  over time, while the frequency signal represents a development of the frequency of the signal filtered by a filter  501 . 
     A schematical setup of filter  501  is illustrated in  FIG. 5   b . Each filter  501  of  FIG. 5   a  may be set up as in  FIG. 5   b , wherein, however, only the frequencies fi supplied to the two input mixers  551  and the adder  552  are different from channel to channel. The mixer output signals are both lowpass filtered by lowpasses  553 , wherein the lowpass signals are different insofar as they were generated by local oscillator frequencies (LO frequencies), which are out of phase by 90°. The upper lowpass filter  553  provides a quadrature signal  554 , while the lower filter  553  provides an in-phase signal  555 . These two signals, i.e. I and Q, are supplied to a coordinate transformer  556  which generates a magnitude phase representation from the rectangular representation. The magnitude signal or amplitude signal, respectively, of  FIG. 5   a  over time is output at an output  557 . The phase signal is supplied to a phase unwrapper  558 . At the output of the element  558 , there is no phase value present any more which is between 0 and 360°, but a phase value which increases linearly. This “unwrapped” phase value is supplied to a phase/frequency converter  559  which may for example be implemented as a simple phase difference former which subtracts a phase of a previous point in time from a phase at a current point in time to obtain a frequency value for the current point in time. This frequency value is added to the constant frequency value fi of the filter channel i to obtain a temporarily varying frequency value at the output  560 . The frequency value at the output  560  has a direct component=fi and an alternating component=the frequency deviation by which a current frequency of the signal in the filter channel deviates from the average frequency fi. 
     Thus, as illustrated in  FIGS. 5   a  and  5   b , the phase vocoder achieves a separation of the spectral information and time information. The spectral information is in the special channel or in the frequency fi which provides the direct portion of the frequency for each channel, while the time information is contained in the frequency deviation or the magnitude over time, respectively. 
       FIG. 5   c  shows a manipulation as it is executed for the bandwidth increase according to the invention, in particular, in the phase vocoder  202   a , and in particular, at the location of the illustrated circuit plotted in dashed lines in  FIG. 5   a.    
     For time scaling, e.g. the amplitude signals A(t) in each channel or the frequency of the signals f(t) in each signal may be decimated or interpolated, respectively. For purposes of transposition, as it is useful for the present invention, an interpolation, i.e. a temporal extension or spreading of the signals A(t) and f(t) is performed to obtain spread signals A′(t) and f′(t), wherein the interpolation is controlled by the spread factor  104 , as it was illustrated in  FIG. 1 . By the interpolation of the phase variation, i.e. the value before the addition of the constant frequency by the adder  552 , the frequency of each individual oscillator  502  in  FIG. 5   a  is not changed. The temporal change of the overall audio signal is slowed down, however, i.e. by the factor  2 . The result is a temporally spread tone having the original pitch, i.e. the original fundamental wave with its harmonics. 
     By performing the signal processing illustrated in  FIG. 5   c , wherein such a processing is executed in every filter band channel in  FIG. 5 , and by the resulting temporal signal then being decimated in the decimator  105  of  FIG. 1 , or in the decimator  205   a  in  FIG. 5   a , respectively, the audio signal is shrunk back to its original duration while all frequencies are doubled simultaneously. This leads to a pitch transposition by the factor  2  wherein, however, an audio signal is obtained which has the same length as the original audio signal, i.e. the same number of samples. 
     As an alternative to the filterband implementation illustrated in  FIG. 5   a , a transformation implementation of a phase vocoder may also be used. Here, the audio signal  100  is fed into an FFT processor, or more generally, into a Short-Time-Fourier-Transformation-Processor  600  as a sequence of time samples. The FFT processor  600  is implemented schematically in  FIG. 6  to perform a time windowing of an audio signal in order to then, by means of an FFT, calculate both a magnitude spectrum and also a phase spectrum, wherein this calculation is performed for successive spectrums which are related to blocks of the audio signal, which are strongly overlapping. 
     In an extreme case, for every new audio signal sample a new spectrum may be calculated, wherein a new spectrum may be calculated also e.g. only for each twentieth new sample. This distance a in samples between two spectrums may be given by a controller  602 . The controller  602  is further implemented to feed an IFFT processor  604  which is implemented to operate in an overlapping operation. In particular, the IFFT processor  604  is implemented such that it performs an inverse short-time Fourier Transformation by performing one IFFT per spectrum based on a magnitude spectrum and a phase spectrum, in order to then perform an overlap add operation, from which the time range results. The overlap add operation eliminates the effects of the analysis window. 
     A spreading of the time signal is achieved by the distance b between two spectrums, as they are processed by the IFFT processor  604 , being greater than the distance a between the spectrums in the generation of the FFT spectrums. The basic idea is to spread the audio signal by the inverse FFTs simply being spaced apart further than the analysis FFTs. As a result, spectral changes in the synthesized audio signal occur more slowly than in the original audio signal. 
     Without a phase rescaling in block  606 , this would, however, lead to frequency artifacts. When, for example, one single frequency bin is considered for which successive phase values by 45° are implemented, this implies that the signal within this filterband increases in the phase with a rate of ⅛ of a cycle, i.e. by 45° per time interval, wherein the time interval here is the time interval between successive FFTs. If now the inverse FFTs are being spaced farther apart from each other, this means that the 45° phase increase occurs across a longer time interval. This means that the frequency of this signal portion was unintentionally reduced. To eliminate this artifact frequency reduction, the phase is resealed by exactly the same factor by which the audio signal was spread in time. The phase of each FFT spectral value is thus increased by the factor b/a, so that this unintentional frequency reduction is eliminated. 
     While in the embodiment illustrated in  FIG. 5   c  the spreading by interpolation of the amplitude/frequency control signals was achieved for one signal oscillator in the filterbank implementation of  FIG. 5   a , the spreading in  FIG. 6  is achieved by the distance between two IFFT spectrums being greater than the distance between two FFT spectrums, i.e. b being greater than a, wherein, however, for an artifact prevention a phase resealing is executed according to b/a. 
     With regard to a detailed description of phase-vocoders reference is made to the following documents: 
     “The phase Vocoder: A tutorial”, Mark Dolson, Computer Music Journal, vol. 10, no. 4, pp. 14-27, 1986, or “New phase Vocoder techniques for pitch-shifting, harmonizing and other exotic effects”, L. Laroche and M. Dolson, Proceedings 1999 IEEE Workshop on applications of signal processing to audio and acoustics, New Paltz, N.Y., Oct. 17-20, 1999, pages 91 to 94; “New approached to transient processing interphase vocoder”, A. Röbel, Proceeding of the 6th international conference on digital audio effects (DAFx-03), London, UK, Sep. 8-11, 2003, pages DAFx-1 to DAFx-6; “Phase-locked Vocoder”, Meller Puckette, Proceedings 1995, IEEE ASSP, Conference on applications of signal processing to audio and acoustics, or U.S. Pat. No. 6,549,884. 
       FIG. 2   b  shows an improvement of the system illustrated in  FIG. 2   a , wherein a transient detector  250  is used which is implemented to determine whether a current temporal operation of the audio signal contains a transient portion. A transient portion consists in the fact that the audio signal changes a lot in total, i.e. that e.g. the energy of the audio signal changes by more than 50% from one temporal portion to the next temporal portion, i.e. increases or decreases. The 50% threshold is only an example, however, and it may also be smaller or greater values. Alternatively, for a transient detection, the change of energy distribution may also be considered, e.g. in the conversion from a vocal to sibilant. 
     If a transient portion of the audio signal is determined, the harmonic transposition is left, and for the transient time range, a switch it a non-harmonic copying operation or a non-harmonic mirroring or some other bandwidth extension algorithm is executed, as it is illustrated at  260 . If it is then again detected that the audio signal is no longer transient, a harmonic transposition is again performed, as illustrated by the elements  102 ,  105  in  FIG. 1 . This is illustrated at  270  in  FIG. 2   b.    
     The output signals of blocks  270  and  260  which arrive offset in time due to the fact that a temporal portion of the audio signal may be either transient or non-transient, are supplied to a combiner  280  which is implemented to provide a bandpass signal over time which may, e.g., be supplied to the tonality correction in block  109   a  in  FIG. 2   a . Alternatively, the combination by block  280  may for example also be performed after the adder  111 . This would mean, however, that for a whole transformation block of the audio signal, a transient characteristic is assumed, or if the filterbank implementation also operates based on blocks, for a whole such block a decision in favor of either transient or non-transient, respectively, is made. 
     As a phase vocoder  202   a ,  202   b ,  202   c , as illustrated in  FIG. 2   a  and explained in more detail in  FIGS. 5 and 6 , generates more artifacts in the processing of transient signal portions than in the processing of non-transient signal portions, a switch is performed to a non-harmonic copying operation or mirroring, as it was illustrated in  FIG. 2   b  at  260 . Alternatively, also a phase reset to the transient may be performed, as it is for example described in the experts publication by Laroche cited above, or in the U.S. Pat. No. 6,549,884. 
     As it has already been indicated, in blocks  109   a ,  109   b , after the generation of the HF portion of the spectrum, a spectral formation and an adjustment to the original measure of noise is performed. The spectral formation may take place, e.g. with the help of scale factors, dB(A)-weighted scale factors or a linear prediction, wherein there is the advantage in the linear prediction that no time/frequency conversion and no subsequent frequency/time conversion is necessitated. 
     The present invention is advantageous insofar that by the use of the phase vocoder, a spectrum with an increasing frequency is further spread and is correctly harmonically continued by the integer spreading. Thus, the result of coarsenesses at the cut-off frequency of the LF range is excluded and interferences by too densely occupied HF portions of the spectrum are prevented. Further, efficient phase vocoder implementations may be used, which and may be done without filterbank patching operations. 
     Alternatively, other methods for signal spreading are available, such as, for example, the PSOLA method (Pitch Synchronous Overlap Add). Pitch Synchronous Overlap Add, in short PSOLA, is a synthesis method in which recordings of speech signals are located in the database. As far as these are periodic signals, the same are provided with information on the fundamental frequency (pitch) and the beginning of each period is marked. In the synthesis, these periods are cut out with a certain environment by means of a window function, and added to the signal to be synthesized at a suitable location: Depending on whether the desired fundamental frequency is higher or lower than that of the database entry, they are combined accordingly denser or less dense than in the original. For adjusting the duration of the audible, periods may be omitted or output in double. This method is also called TD-PSOLA, wherein TD stands for time domain and emphasizes that the methods operate in the time domain. A further development is the MultiBand Resynthesis OverLap Add method, in short MBROLA. Here the segments in the database are brought to a uniform fundamental frequency by a pre-processing and the phase position of the harmonic is normalized. By this, in the synthesis of a transition from a segment to the next, less perceptive interferences result and the achieved speech quality is higher. 
     In a further alternative, the audio signal is already bandpass filtered before spreading, so that the signal after spreading and decimation already contains the desired portions and the subsequent bandpass filtering may be omitted. In this case, the bandpass filter is set so that the portion of the audio signal which would have been filtered out after bandwidth extension is still contained in the output signal of the bandpass filter. The bandpass filter thus contains a frequency range which is not contained in the audio signal  106  after spreading and decimation. The signal with this frequency range is the desired signal forming the synthesized high-frequency signal. In this embodiment, the distorter  109  will not distort a bandpass signal, but a spread and decimated signal derived from a bandpass filtered audio signal. 
     It is further to be noted, that the spread signal may also be helpful in the frequency range of the original signal, e.g. by mixing the original signal and spread signal, thus no “strict” passband is necessitated. The spread signal may then well be mixed with the original signal in the frequency band in which it overlaps with the original signal regarding frequency, to modify the characteristic of the original signal in the overlapping range. 
     It is further to be noted that the functionalities of distorting  109  and filtering  107  may be implemented in one single filter block or in two cascaded separate filters. As distorting takes place depending on the signal, the amplitude characteristic of this filter block will be variable. Its frequency characteristic is, however, independent of the signal. 
     Depending on the implementation, as illustrated in  FIG. 1 , first the overall audio signal may be spread, decimated, and then filtered, wherein filtering corresponds to the operations of the elements  107 ,  109 . Distorting is thus executed after or simultaneously to filtering, wherein for this purpose a combined filter/distorter block in the form of a digital filter is suitable. Alternatively, before the (bandpass-) filtering ( 107 ) a distortion may take place here when two different filter elements are used. 
     Again, alternatively, a bandpass filtering may take place before spreading so that only the distortion ( 109 ) follows after the decimation. For these functions two different elements are of advantage here. 
     Again alternatively, also in all variants above, the distortion may take place after the combination of the synthesis signal with the original audio signal such as, for example, with a filter which has no, or only very little effect, on the signal to be filtered in the frequency range of the original filter, which, however, generates the desired envelope in the extended frequency range. In this case, again two different elements may be used for extraction and distortion. 
     The inventive concept is suitable for all audio applications in which the full bandwidth is not available. In the propagation of audio contents such as, for example, by digital radio, Internet streaming and in audio communication applications, the inventive concept may be used. 
     Depending on the circumstances, the inventive method may be implemented for analyzing an information signal in hardware or in software. The implementation may be executed on a digital storage medium, in particular a floppy disc or a CD, having electronically readable control signals stored thereon, which may cooperate with the programmable computer system, such that the method is performed. Generally, the invention thus consists in a computer program product with a program code for executing the method stored on a machine-readable carrier, when the computer program product is executed on a computer. In other words, the invention may thus be realized as a computer program having a program code for performing the method, when the computer program is executed on a computer. 
     While this invention has been described in terms of several embodiments, there are alterations, permutations, and equivalents which fall within the scope of this invention. It should also be noted that there are many alternative ways of implementing the methods and compositions of the present invention. It is therefore intended that the following appended claims be interpreted as including all such alterations, permutations, and equivalents as fall within the true spirit and scope of the present invention.