Patent Publication Number: US-6223313-B1

Title: Method and apparatus for controlling and observing data in a logic block-based asic

Description:
FIELD OF THE INVENTION 
     The invention generally relates to application specific integrated circuits (ASICs), and more specifically, the invention relates to testing gate arrays. 
     BACKGROUND OF THE INVENTION 
     Use of ASICs has become widespread in the semiconductor industry as giving circuit design engineers a relatively high amount of circuit functionality in a relatively small package. In particular, ASICs are customizable integrated circuits that are customized to implement a circuit specified by a design engineer (a “user-designed circuit”). One type of ASIC is the gate array, which generally includes an array of function blocks, each of which are predesigned and/or prefabricated to include a particular number, arrangement, and type of semiconductor devices, e.g., transistors. To customize a gate array to implement a particular user-designed circuit specified by a design engineer, various connections are made among the semiconductor devices within the function block and/or various connections are made among function blocks (i.e., routing is customized). 
     ASICs, including gate arrays, once customized to implement a user-designed circuit, must be tested to ensure that the user-designed circuit operates properly. Tests must be able to detect faults, which are the results of defects (physical problems with the circuit, e.g., shorts, and/or improper circuit design), resulting in improper or unexpected circuit behavior. 
     Faults include “Stuck At Faults” (SAFs), delay faults and current faults. SAFs occur when a particular connection in the circuit remains at (is “stuck” at) a logical low level or a logical high level regardless of what signals are applied to the circuit. (As used herein, “logical low” refers to a “0” signal, which is often a ground signal. A “logical high” refers to a “1” signal, which is often a V DD  signal.) Delay faults occur when the circuit is designed to accommodate a particular propagation time, but the circuit actually operates much slower than expected. For instance, if a circuit was designed with the belief that there would only be a 5 ns propagation time of a signal between a first point and a second point, but in operation the signal actually takes 15 ns to propagate from the first point to the second point, the circuit may not operate properly. Current faults will often occur in circuits utilizing CMOS devices, which are not supposed to draw current when inactive, but do. The defects that cause current faults can frequently be detected by testing for SAFs. 
     Well-designed tests of an integrated circuit will generally be able to detect most SAFs at the gate level (i.e., the conceptual circuit design level containing Boolean logic, flip-flops, etc.) by testing all connections between logic elements. In order to test all connections between logic elements, the tester needs to be able to both (1) control, or set, the value at a particular connection and (2) be able to observe the value at the particular connection. For instance, in order to test the connection between point A and point B for Stuck At 0 faults, the tester needs to be able to apply stimulus data that ought to place a logical high on the connection line, and then the tester needs to be able to observe the connection to see if and how the value changes as a result of the stimulus data. 
     In discrete circuit design, the ability to control and observe a particular circuit is often done by simply probing the various connections among the logic elements of the circuit. However, with integrated circuits, the ability to probe connections internal to the circuit is generally unavailable and other methods of testing have had to be developed as a result. 
     One method of testing an integrated circuit that enjoys the most popularity among IC designers is “scan” testing, which will be described with reference to the block diagrams of FIGS. 1 and 1 a.  In FIG. 1, circuit  102  is generally composed of any number and arrangement of logic elements (e.g., Boolean logic gates, flip-flops, latches, etc.) and has input A and output B. Inputs A can be coupled directly to flip-flops  104  (via lines  114 ) or to other logic elements in logic  102 . Likewise, outputs from flip-flops  104  can be coupled directly to outputs B (via lines  112 ), to other logic elements in logic  102 , or directly to other flip-flop  104  inputs. Each flip-flop  104  contained in circuit  102  is coupled to a clock signal such as CLK 1   108  or CLK 2   109 . The flip-flops  104 , shown apart from the circuit  102  for illustrative purposes only, will each, upon receiving a triggering clock edge, store a value and hold the value on its respective output until a next triggering clock edge is received. Therefore the flip-flops of circuit  102  collectively represent the state of the circuit: at any time when the clocks are stopped, the flip-flops will maintain the state of the circuit. 
     By taking advantage of the state-machine nature of the circuit, the state of the circuit  102  can be controlled for test purposes by placing known values into the flip-flops  104 . Similarly, the state of the circuit can also be observed by reading the values held in the flip-flops after the circuit has been run. In order to control and observe the values held in flip-flops  104  of the circuit  102 , the flip-flops  104  are, in addition to their regular circuit connections represented by lines  112  and  114 , coupled to one another in a daisy-chain fashion, i.e., the output of one flip-flop is coupled to the input of the next flip-flop, as generally shown in FIG. 1 a.  Furthermore, clock steering logic, such as multiplexer  111 , is frequently inserted so that all testing and shifting can be effected with one clock signal. To test logic circuit  102 , the regular “mission mode” operation of circuit  102  is stopped and a series of stimulus values are shifted into flip-flops  104 , via the daisy-chain, so that each flip-flop in logic circuit  102  has a known value. Stimulus values are shifted in by applying the stimulus values one at a time to the input  106  of the first flip-flop in the daisy-chain and running the circuit clock  108  (coupled to the clock input of each flip-flop  104 ) to propagate the values through the daisy-chain. After the flip-flops  104  have each received a known test value, the circuit  102  is then exercised (run normally) for a brief period, e.g., one clock cycle, and then stopped. The state of the circuit resulting from its being run is captured in flip-flops  104 . The resulting values are then shifted out of the flip-flops  104 , by again running the clock  108  and reading the values at the output  110  of the last flip-flop in the daisy-chain. 
     More specifically, to implement scan-type testing, typically one of two techniques is used: mux-based scan or clock-based scan. “Mux-based scan” is the more commonly used technique and is described with reference to the block diagram of FIG.  2 . Clock-based scan will be described with reference to FIG.  3 . 
     As shown in FIG. 2, for each flip-flop  104   n  in the logic circuit  102 , (where flip-flops  104  are shown apart from circuit  102  for illustrative purposes only) a 2-input multiplexor  212   n  is placed at the D-input of each respective flip-flop  104   n . One input, e.g., the 0 input, for each multiplexor  212   n  receives the regular connection  114  from the logic  102  that would otherwise go directly into the D-input but for the multiplexor  212   n . The second input, e.g., the 1 input, of each multiplexor  212   n  is coupled to the output of a flip-flop  104   n+1 , thereby daisy-chaining the flip-flops. As shown in FIG. 2, the Q-output of flip-flop  104   2  is coupled to the 1-input of multiplexor  212   1 , and the Q-output of flip-flop  104   1  would be coupled to another multiplexor  212   0  (not shown). The 1-input to multiplexor  212   2  would be received from the Q-output of flip-flop  104   3  (not shown). A circuit clock line (CLK)  108  is coupled to each of the flip-flops  104   n  as it would be without inclusion of multiplexors  212   n . A SHIFT signal  214  is coupled to the select input of each of the multiplexors  212   n . When SHIFT  214  is a logical low, the circuit operates normally, as if the multiplexors were not present. Such normal circuit operation can be used for regular mission mode operation as well as for exercising circuitry during test modes. When SHIFT is a logical high, the circuit is placed in a “shift mode” of operation and test data (stimulus or result values) is shifted into or out of flip-flops  104   n  by application of a clock signal on CLK  108 . 
     In FIG. 2, to test circuit  102 , the circuit  102  is first placed in shift mode by placing a logical high signal on SHIFT  214 . The clock signal on CLK  108  is run in a controlled manner while stimulus values are shifted into the flip-flops  104 . Once stimulus values are in place, SHIFT  214  is brought to a logical low. The clock signal on CLK  108  is run but controlled to ensure that only a limited number of clock cycles are run, e.g., one clock cycle. Resulting values are then captured in the flip-flops  104   n  by operation of the last clock edge in this test sequence, also sometimes referred to as a “capture clock.” SHIFT is then brought to a logical high, re-entering shift mode, and a signal on CLK  108  is applied to allow the captured data to be shifted out of the flip-flops  104   n . 
     While the above testing method is useful as described for detecting SAFs, mux-based scan can also be used to test for delay faults. To do so, test data would be shifted into the flip-flops  104   n  as described above. Then two clock edges—“a launch clock” and “a capture clock”—are applied on CLK  108  with controlled timing between them. The “launch clock” is the clock edge that places the circuit in a state ready for test. In delay fault testing, the first clock edge that occurs after the stimulus data is finally positioned in flip-flops  104  is the launch clock, while in the SAF testing scenario described above, the launch clock would essentially be the last clock edge to occur in shift mode. The “capture clock” is the clock edge at which resulting values are captured in the flip-flops  104 , and is similar in both delay fault and SAF testing. After the launch clock and the capture clock have been applied, data is shifted out of the flip-flops  104   n  as described above. If the captured data does not correspond to that expected, then a delay fault may be detected. 
     The flip-flops already present in the user&#39;s circuit design are used for testing when using a mux-based scan technique, although each flip-flop in logic circuit  102  is essentially replaced with a multiplexor/flip-flop combination to enable daisy-chaining. Nonetheless, the flip-flops present in the user&#39;s logic circuit  102  will often be insufficient to fully test the circuit  102  and additional flip-flop/multiplexor circuitry will need to be added at various points in the design to achieve adequate fault coverage. 
     Moreover, there are many “design-for-test” rules (DFT rules) that have become generally known and used in designing user-designed circuits as a direct consequence of mux-based scan in order to avoid problems during testing. These DFT rules include the following: 
     Circuits should preferably not be designed to include falling-edge triggered flip-flops. Otherwise, some flip-flops would be clocked on the rising edge of the circuit clock, and some would be clocked on the falling edge. In such a situation, during a test data shift in, some of the flip-flops may not receive appropriate stimulus values and to avoid this situation extra test flip-flops may need to be included in the daisy-chain. 
     Clocks should only be designed to be coupled to clock pins and not to the D-input of a flip-flop or a gate that ultimately is coupled to the D-input of a flip-flop. Otherwise, setup and hold time violations may occur during test mode and the circuit will not reliably capture response values. 
     The Q-output of a flip-flop should not be directly or indirectly (e.g., through combinational logic or drivers) coupled to the clock input of another flip-flop (such as in a Johnson counter), as that clock-input will not be adequately controllable during testing. More generally, clock inputs throughout the circuit must be controllable for testing the circuit. 
     All gates through which the clock passes must also be controlled during testing to allow the clock to pass uninfluenced by other values during test value shifting. For instance, if the clock signal is applied to the first input of a 2-input AND-gate, where the AND-gate output is applied to the clock input of a flip-flop, then the second input to the AND-gate must be held to a logical high during a test value shift. 
     All asynchronous clear and reset pins must be gated so they can be prevented from interfering with shift mode. 
     Other DFT rules are also commonly known. Many of these DFT rules are a direct result of the fact that testing, e.g., controlling and observing values, can only be done with static patterns (logical high and logical low values)—clock edge transitions can not be generated. 
     Thus, not only does mux-based scan add additional circuitry (e.g., muxes and additional flip-flops) to the circuit, taking up considerable real estate on an integrated circuit, but the DFT rules, which have often developed as a result of the limitations of mux-based scan, have placed considerable limits on the design of the circuit, all to simply allow testing of the circuit. 
     A second type of scan technique is “clock-based scan”, described with reference to the block diagram of FIG.  3 . Rather than replacing each flip-flop in the logic circuit  102  with a mux/flip-flop combination as in mux-based scan, the flip-flops  104   n  in the logic design  102  are replaced with a dual interface flip-flop  304  shown in FIG.  3 . Flip-flop  304  is composed of one flip-flop having two interfaces: one interface is shown in lower portion  310  and one interface is shown in the upper portion  312  of flip-flop  304 . When placed in a circuit  102 , the inputs and outputs (D, Q, CLK) of lower interface  310  are coupled to receive signals used for normal operation (mission mode). The inputs and outputs (TD, TQ) of upper interface  312  are coupled with the upper interface of other flip-flops  304  to form a daisy-chain, and TCLK is coupled to receive a test clock signal, which can be distinct from the regular circuit clock (the “user clock”) coupled to CLK. 
     A signal input to SHIFT  314  indicates whether the upper interface or the lower interface should be active. When the signal coupled to the SHIFT input  314  is a logical low, the upper interface  312  may be inactive while the lower interface  310  must be active. Thus, when the signal on SHIFT  314  is low, the circuit  102  behaves in mission mode. When the signal on SHIFT  314  is a logical high, the lower interface  310  must be inactive and the upper interface  312  must be active. Stimulus values are shifted into the respective flip-flops  304  via the daisy chained upper-interfaces  312 , using TCLK to control the shift mode. Once stimulus values are in place, the circuit will be run in test mode for a controlled time period (i.e., SHIFT receives a logical low), after which SHIFT is again asserted high to enable captured values to be shifted out under control of TCLK. As will be understood by those of skill in the art, clock-based scan can easily mimic mux-based scan. As is also known in the art, device  304  may be a latch having two interfaces (one for mission and test modes and one for shift mode) rather than a flip-flop. 
     Clock-based scan is advantageous over mux-based scan in that clock-based scan has fewer DFT rules associated with it. Since a separate interface and test clock are used for testing, most clock related DFT rules will no longer need to be followed when designing the underlying circuit. Nonetheless, clock-based scan tends to be more expensive than mux-based scan, causing it to be used less frequently than mux-based scan. In addition, even in clock-based scan, flip-flops will need to be added into the user-designed circuit in places where those present in the user-designed circuit are insufficient for test purposes. 
     Thus, although scan techniques are widely used to test integrated circuits, they are replete with limitations. Not only do scan techniques add a considerable number of devices to the design of the circuit (e.g., muxes and/or flip-flops), taking up considerable real estate, but use of such additional circuitry will limit the availability of circuitry that could otherwise be used for regular circuit operation. In other words, in a gate array, to insert a multiplexor and/or flip-flop into the circuit for test purposes, various function blocks, or portions thereof, will be sacrificed for testing purposes and become unusable for implementing the logic to be used for the regular (non-test) portion of the circuit. Moreover, when using scan techniques, and mux-based scan in particular, numerous DFT rules will need to be followed just to allow testing of the circuit, limiting the flexibility of circuit design or considerable additional expense will be required to include appropriate dual-interface devices for clock-based scan. 
     To make it even more difficult to design integrated circuits, test circuitry cannot be inserted into the circuit until the circuit design is substantially complete. More specifically, the knowledge of where the circuit&#39;s flip-flops and/or latches are located in the circuit and how they are connected is critical to the ability to insert additional appropriate test circuitry. Typically, once the circuit is designed, then the design is modified for testing purposes, replacing flip-flops in the circuit with a multiplexor/flip-flop combination or a dual-interface flip-flop. Then the circuit must be further analyzed to determine if portions of the circuit need to have additional flip-flops or clock steering logic inserted to adequately test the circuit. Thus, test circuitry can only be added with prior knowledge on the part of the IC designer of the completed circuit design. If space is an issue on the IC, the user&#39;s circuit may need to be redesigned to accommodate the test structures. 
     In addition, the nature of scan requires that once stimulus data is placed in the respective flip-flops, the entire circuit must be exercised simultaneously. Hence, considerable computational effort will have to be made in designing test vectors in order to ensure that appropriate stimulus values will be placed in the circuit as well as to ensure that all parts of the circuit that can influence resulting captured values have been accounted for. Further, having to exercise the entire circuit simultaneously makes faults more difficult to isolate as the source of the problem may not be immediately apparent. 
     Therefore, a gate array design that inexpensively (in terms of real estate and other resources) implements the testing of a circuit implemented by the gate array without prior knowledge of the circuit, permits full testing of the circuit without complex DFT rules, and allows portions of the circuit to be exercised in isolation from the rest of the circuit, saving considerable time and computational effort, would represent an advancement in the art. 
     SUMMARY OF THE INVENTION 
     A system for testing an integrated circuit, and particularly a gate array, is disclosed which overcomes the deficiencies and limitations discussed above. In general, a system in accordance with the invention includes an array of logic blocks, where the logic blocks include a predesigned arrangement of devices and where the logic blocks are couplable to form a user-designed circuit, e.g., by customizing routing. Within the logic blocks and prior to any knowledge of the user-designed circuit, the predesigned arrangement of devices includes logic to enable testing of the user-designed circuit to be later formed. The logic to enable testing allows the logic blocks to be selectable to operate in a freeze mode, a mode unlike the modes of conventional scan, or to operate in normal mode. When selected to operate in a normal mode of operation, each of the logic blocks forms part of the user-designed circuit (when customized), and may operate to perform a sequential logic function, a combinational logic function, a memory logic function, or other function. When selected to operate in freeze mode, the logic blocks operate together as a series of daisy-chained master-slave flip-flops, nonresponsive to user signals, regardless of the underlying function performed by each of the logic blocks in its respective normal mode of operation, as dictated by the user-designed circuit. 
     Also advantageous over conventional scan, the logic blocks in the array are further predesigned so that each logic block is equipped with addressable mode control, allowing each logic block to be individually selected to exercise (operate in normal mode) while other logic blocks simultaneously remain frozen, driving out stimulus data values. 
     To test a user-designed circuit, all the logic blocks are first selected, via addressable mode control, to be frozen, thereby forming daisy-chained flip-flops. Stimulus values are shifted into the flip-flops via the daisy-chain. Once shifting has completed, certain isolated and/or nonadjacent logic blocks, selected by addressable mode control, are exercised (i.e., permitted to operate in a normal mode of operation) to verify their operation. Thereafter, all logic blocks are again frozen and then captured data is shifted out of the array. Such a method of testing the user-designed circuit is referred to herein as partition test. Moreover, as discussed previously, such a method of isolated testing is not generally achievable with conventional techniques. 
     Another method of testing the operation of a user-designed circuit in accordance with the invention is “scan-emulation” testing. To perform scan-emulation testing, first all logic blocks are frozen, via addressable mode control, and then stimulus values are shifted into the logic blocks via the daisy-chained flip-flops. Then all logic blocks are selected, also via addressable mode control, to operate in normal mode. Thereafter, all logic blocks are again selected to freeze. Data captured in the frozen logic blocks (i.e., in the daisy-chained flip-flops) is then shifted out of the array. 
     During shifting, in addition to the logic blocks each behaving as a flip-flop, the master latch and the slave latch of each flip-flop are capable of latching independent data values, allowing independent control and observation of each latch in the respective flip-flops. To do so, the test clock is independently enabled and disabled to each latch. 
     Implementing predesigned logic blocks in accordance with the invention permits fully testing a gate array without the user (IC designer) having to insert test structures into the user-designed circuit unlike in conventional scan techniques. Moreover, circuitry to enable testing in accordance with the invention is placed in logic blocks prior to any knowledge of the user-designed circuit, and therefore testing of the user-designed circuit does not depend on placement of flip-flops or other sequential logic in the user-designed circuit. 
     In addition, a logic block in accordance with the invention is used for dual purposes: (1) during test, it can be selected in an addressable manner to freeze, driving out stimulus data, or (2) the same circuitry that can drive out stimulus data, when not selected to be frozen, will perform in normal mode whatever function is dictated by the user-designed circuit, including combinational, sequential, or other functions. Thus, only a minimal number of devices need be included in the predesigned logic blocks prior to customization by a user to enable test. 
     Further, logic blocks predesigned in accordance with the invention impose few limitations on designers of user-designed circuits in the form of DFT rules. Thus, using an array including logic blocks in accordance with the invention will have high flexibility for the user in terms of user circuit design. Moreover, because the logic blocks are equipped with addressable mode control, small portions of a user-designed circuit can be tested in isolation. Still, a scan-emulation method of testing can be used with logic blocks predesigned in accordance with the invention for those design engineers who prefer that technique. 
     More specific details of test circuitry in accordance with the invention are described below. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention will be described with respect to particular embodiments thereof, and reference will be made to the drawings, which are not necessarily drawn to scale, and in which: 
     FIG. 1 is a generalized functional block diagram of a circuit including a plurality of flip-flops; 
     FIG. 1 a  shows the diagram of FIG.  1  and generally illustrates scan techniques; 
     FIG. 2 is a more detailed representation of FIG. 1 when mux-based scan is utilized; 
     FIG. 3 is a functional block diagram of a device used in clock-based scan; 
     FIG. 4 is a generalized block diagram of an ASIC in accordance with one embodiment of the invention; 
     FIG. 5 is a generalized block diagram of a function block  420  in accordance with one embodiment of the invention; 
     FIG. 6 is a functional block diagram of a stage two logic block  444  in accordance with one embodiment of the invention not showing the circuitry implementing test; 
     FIG. 7 is a schematic diagram of FIG. 6; 
     FIG. 8 is a functional block diagram of combinational logic that can be implemented by the circuit shown in FIG. 6; 
     FIG. 9 is a functional block diagram of a D-type flip-flop; 
     FIG. 10 is a more detailed functional block diagram of the D-type flip-flop of FIG. 9, including a master latch and a slave latch; 
     FIG. 11 is the functional block diagram of FIG. 6 further including circuitry to enable testing in accordance with the invention; 
     FIG. 12 is a schematic diagram of FIG. 11; 
     FIG. 13 is a more generalized functional block diagram of the circuit of FIG. 11; 
     FIG. 14 is a functional block diagram of an array in accordance with the invention; 
     FIG. 15 is a block diagram representation of an array in accordance with the invention when using partition test; 
     FIG. 16 is a block diagram showing that during scan emulation testing, extra bits of data, that would not be present in traditional scan techniques, may be present when using an array in accordance with the invention; 
     FIG. 17 is a block diagram of an STLB broken into master and slave latch; 
     FIG. 18 is a functional block diagram of an array, including test support circuitry, in accordance with the invention; 
     FIG. 19 is a functional block diagram of a tri-state driver stage  462  in accordance with the invention; 
     FIG. 20 is a schematic drawing of the circuit shown in FIG. 19; 
     FIG. 21 is a block diagram illustrating potential contention on tri-state driver outputs during shift mode; and 
     FIG. 22 is the circuit shown in FIG. 20 with additional circuitry added to prevent tri-state contention during shift. 
     FIG. 23 is a block diagram illustrating “adjacent” and “nonadjacent” STLBs. 
    
    
     DETAILED DESCRIPTION 
     A generalized block diagram of an ASIC  400  in accordance with the invention is shown in FIG.  4 . ASIC  400  includes an array  410  of function blocks  420 . In one embodiment of the invention, each function block  420  is identical to the other function blocks in array  410 , although other embodiments of the invention allow for variance among function blocks. Some embodiments may include one or more regions  421 , which contain circuitry such as memory blocks or logic cores. Also shown in FIG. 4 is periphery area  430  surrounding array  410 . Periphery area  430  includes circuitry such as IO pads, test support circuitry, and other support circuitry for array  410 . 
     Each function block  420  can be configured to perform combinational functions, sequential functions and/or memory functions (e.g., SRAM). As shown in FIG. 5, one embodiment of function block  420  is composed of three modules: two computation modules  440 . 1  and  440 . 2  and a communication module  460 . Each module has a predesigned internal architecture, including a predesigned arrangement and coupling of semiconductor devices, but the function performed by each module can be varied by varying input signals to each respective module. Input signals may be varied by coupling the input to a logical high signal, a logical low signal, the output of the same or different module, or a signal from off-chip. 
     Computation modules  440 . 1  and  440 . 2  are identical mirror images of each other in one embodiment of the invention and are generally referred to with reference number  440 . Computation module  440  includes a “stage one” logic block  442  and “stage two” logic block  444 . In one embodiment of the invention, stage one logic block  442  is a four-input multiplexor (not shown). Stage two logic block (STLB)  444  will be described in more detail with respect to FIG.  6 . 
     Communication module  460  includes a driver stage  462  and two identical inverter stages  464  in one embodiment of the invention. Each inverter stage  464  contains at least one inverter (not shown). The driver stage  462  will be described in more detail with respect to FIG.  19 . 
     A more detailed description of one embodiment of the internal circuit structure of each function block  420 , including computation and communication modules  440 . 1 ,  440 . 2 , and  460 , can be found in application Serial No. 08/821,475, filed on Mar. 21, 1997, now U.S. Pat. No. 6,014,038, entitled “Function Block Architecture for Gate Array,” and incorporated by reference herein. 
     STLBs 
     An embodiment of stage two logic block (STLB)  444  of computation module  440  is functionally shown in FIG.  6 . STLB  444  generally includes three multiplexors  660 ,  680 , and  696  and two bit storage units  670  and  688 . First multiplexor  660  has a first input for receiving a signal DS on line  662  and a second input which is coupled to the output of bit storage unit  688  via line  694 . DS serves as a signal input into STLB  444 . Multiplexor  660  has two internal paths to its output on line  664 . The first, or upper, path couples the input on line  694  to multiplexor output on line  664  when switch  663  is closed. Switch  663  is controlled by signal F on line  668 . The second, or lower, path couples the DS signal on line  662  to the output on line  664  when switch  661  is closed. Switch  661  is controlled by signal M on line  666 . Multiplexor  660 , and others like it, are herein referred to as “dual-control multiplexors.” 
     Bit storage unit  670  receives as an input the output from dual-control multiplexor  660  on line  664 . In one embodiment of the invention, bit storage unit  670  is a pair of cross-coupled inverters  672  and  674  as shown in FIG.  6 . Inverter  674  is generally designed to be weaker than inverter  690  in order to allow any changing bit outputs from multiplexor  660  to be placed in bit storage unit  670  by overdriving inverter  674 . In addition, inverter  674  is enabled and disabled by signal M on line  666 . Thus, bit storage unit  670  can be configured to appear as a simple inverter in certain configurations of computation module  440 . 
     Dual-control multiplexor  680  receives as a first input the output of bit storage unit  670  on line  676 . The other input to multiplexor  680  is coupled to signal DA on line  678 , an input into STLB  444 . Similar to multiplexor  660 , multiplexor  680  has two signal paths, each controlled by a respective switch  681  or  683 . Signal S on line  682  controls switch  681  while signal L on line  684  controls switch  683 . 
     Bit storage unit  688  receives as an input the output of dual-control multiplexor  680  on line  686 . Like bit storage unit  670 , bit storage unit  688  is, in one embodiment, composed of a pair of cross-coupled inverters  690  and  692 , where inverter  692  is weaker than inverter  672 , and where inverter  692  is selectively enabled by signal L on line  684 . 
     Multiplexor  696  receives as a first input the output of bit storage unit  688  on line  694 . The second input to multiplexor  696  is received from the output  676  of bit storage unit  670 . Multiplexor  696  further has a select input SX, which multiplexor  696  receives on line  697  and which selects one of the multiplexor&#39;s inputs to be output onto line  698 . 
     Line  698  is coupled to inverter  700 , which serves as a buffering mechanism and which outputs signal Q on line  702 . 
     In addition, STLB  444  also includes select and enable logic, which selects the various switches in multiplexors  660  and  680  as well as enables inverters  674  and  692  in bit storage units  670  and  688 , respectively. The select and enable logic in one embodiment includes NOR gate  704 , NAND gate  710 , NAND gate  716 , and inverters  722  and  724 . 
     NOR gate  704  has a first input MC on line  706 , an input into STLB  444 , and a second input received from the output of NAND gate  710  via line  668 . NOR gate  704  outputs signal M on line  666 , which controls switch  661  and enables inverter  674 . 
     Inputs to NAND gate  710  are EN on line  712  and AS on line  714 , both inputs to STLB  444 . The output  668  from NAND gate  710  is the signal F which controls switch  663  in multiplexor  660 . 
     NAND gate  716  receives as inputs signal SC on line  718  and signal S 2  on line  720 , both inputs to STLB  444 . The output of NAND gate  716  is coupled to inverter  724 , which outputs signal S on line  682  to control switch  681  in multiplexor  680 . 
     Inverter  722  also receives signal S 2  on line  720  and outputs signal L on line  684  to control switch  683  of multiplexor  680  as well as inverter  692  in bit storage unit  688 . 
     While select and enable logic for STLB  444  is shown in FIG. 6 as NOR, NAND, and inverting gates, a person of ordinary skill in the art will recognize that a number of other gate combinations are possible. Further, select and enable logic is not shown fully connected in FIG. 6 to aid in the clarity of the figures. However, the connections should be clear to those of skill in the art by the signal names and/or line reference numbers provided. Similar techniques are employed for clarity in other figures as well. 
     FIG. 7 shows one specific implementation of the functional model shown in FIG.  6 . Transmission gates are used to implement the multiplexor/switch functions shown in FIG.  6 . Pass gates would also be an acceptable alternative to transmission gates in other embodiments of the invention. Specific details of FIG. 7 will not be further discussed as they will be clear to one of skill in the art upon a comparison of FIG. 7 with FIG.  6 . It should further be clear to one of ordinary skill in the art that, although logic gates and inverters are shown, the logic gates in FIG.  7  and other figures are implemented with various transistor configurations. Moreover, one skilled in the art will recognize in FIG.  7  and other figures that various additions, deletions and/or rearrangements of inverters and/or logic gates will still result in an equivalent circuit. 
     As shown in FIG. 6, STLB  444  is quite flexible and can, by varying the signals coupled to its inputs, be configured to operate as a combinational, sequential, or memory function, as described in detail in U.S. application Ser. No. 08/821,475, now U.S. Pat. No. 6,014,038, incorporated by reference above. For instance, STLB  444  can implement a combinational function, such as a 2-input AND gate whose output is coupled to one of the inputs of a 2-input NOR gate, as shown in FIG.  8 . To do so, signal A is coupled to be received on input SX. Signal B is coupled to be received on input EN and signal C is coupled to be received on input AS, while DA, DS, MC, SC, and S 2  are all tied to a logical low signal. Output Q on line  702  is the output of the function. 
     The circuit shown in FIG. 6 can also be configured to implement a sequential function. FIG. 9 shows a flip-flop  730  including D-input  732 , clock input  734 , and Q output  736 . FIG. 10 shows the flip-flop of FIG. 9 broken down into a master latch  740  and a slave latch  750 . In FIG. 10, the Q output of the master latch  740  is coupled by line  742  to the D input of the slave latch  750 . The slave latch  750  produces Q output  736 . To implement the flip-flop shown in FIGS. 9 and 10, the circuit of FIG. 6 is configured with its inputs in the following manner. The data input (D) to the flip-flop is coupled to input DS. The output Q from STIB  444  is also the Q output of the flip-flop shown in FIG. 9. A clock signal is applied to input MC and SC. Additionally, if enable, clear and/or preset are desired, an enable signal is applied to input EN and a clear signal or a preset signal is applied to inputs AS and S 2 . DA is connected to a logical low for a clear signal or to a logical high for a preset signal. SX is tied to a logical low. In this manner, a data bit input at DS will pass serially through bit storage unit  670  and bit storage unit  688 , and the implementation acts as a master-slave flip-flop similar to that shown in FIG.  10 . 
     When bit storage units  670  and  688  are implemented with cross-coupled inverters in an embodiment of the invention, inverters  672  and  690  should be stronger than inverters  692  and  674 , respectively. Thus, when the input data from the multiplexor to the bit storage unit changes states (e.g., 0 to 1), the input data will overdrive inverter  674  and/or inverter  692 . When switch  661  opens (on a high clock signal), the cross-coupled inverters of bit storage unit  670  remain undisturbed and hold the last bit stored. In like manner, when switch  681  opens, the cross-coupled inverters of bit storage unit  688  remain undisturbed and hold the last bit stored. 
     A latch can be implemented in a similar manner as a flip-flop, but only one bit storage unit needs to be utilized. To form a latch with an active low gate, S 2  would serve as the latch gate, while the DA line would serve as the D input to the latch and Q would serve as the latch output. SC, MC, DS, and SX would all be held to a logical low while AS and EN would be held at a logical high. To form a latch with an active high gate, SC would serve as the latch gate, DS would serve as the D input to the latch, and Q would be the latch output. S 2 , EN, and AS would be held to a logical high while MC, SX, and DA would each be held to a logical low. 
     STLB Modified for Test 
     According to traditional thinking with respect to testing integrated circuits, the ability to get good fault coverage in testing a circuit is usually contingent upon insertion of scan flip-flops within the circuit to set and/or store the state of the circuit under test, as previously discussed. Multiplexors must be inserted in front of flip-flops already present in the design as discussed with reference to FIG. 2 or flip-flops must be converted to the dual-interface devices shown and discussed with respect to FIG.  3 . If flip-flops already present in the circuit are not adequate to fully test the circuit, additional circuitry (e.g., multiplexors and/or flip-flops) must be added to the circuit solely for test purposes. Thus, when an IC designer desires to implement certain designs in a gate array, some circuitry (e.g., function modules or portions thereof) in the gate array will have to be sacrificed for testing purposes. 
     In order to test the circuit of FIG. 6 as shown, traditional test structures would need to be utilized. In other words, to use mux-based scan for every STLB  444  in array  410  that implements a flip-flop, other logic in the same or other function blocks  420  will have to be coupled to implement a multiplexor whose output is coupled to the respective flip-flop (e.g., the multiplexor included in one embodiment of stage one logic  442  could be used for such a purpose). To use clock-based scan would be even more complicated. Having to use certain logic to implement structures for testing purposes typically eliminates the ability to use that same circuitry for other purposes. 
     Nonetheless, a gate array predesigned in accordance with the invention minimizes the sacrifice of all or part of the circuitry of any function block  420  strictly for testing purposes and does so in a way that avoids having to design a user-designed circuit in accordance with conventional DFT rules as described previously. Further, a predesigned gate array in accordance with the invention allows testing of a user-designed circuit without regard to where sequential logic will be placed in the user-designed circuit to be ultimately implemented by the gate array. 
     In accordance with the invention, although not always ultimately configured to perform a sequential function in normal operation, because the STLB  444  of FIG. 6 can easily be configured to perform sequential functions, i.e., since it includes the ability to store bits, circuitry for enabling testing of the array is placed in each STLB  444 . Use of each STLB  444  for testing purposes can actually occur regardless of the underlying type of user-defined circuit functionality (e.g., combinational, sequential) the STLB  444  implements once it has been customized (i.e., by coupling its inputs to various signals). Hence, testing of a circuit implemented by the array in accordance with the invention does not require prior knowledge of the circuit nor involvement of the designer. 
     FIG. 11 shows FIG. 6 as modified to enable testing in one embodiment of the invention. As shown in FIG. 11, two additional dual-control multiplexors  802  and  804  are added to the circuit of FIG.  6 . The output of dual-control multiplexor  660  is coupled to the first input of dual-control multiplexor  802  while the second input of dual-control multiplexor  802  is coupled to receive test data (TD) on line  806 . The output of dual-control multiplexor  802  is coupled to bit storage unit  670 . Dual-control multiplexor  804  has its first input coupled to the output of dual-control multiplexor  680  and its second input coupled to the output of bit storage unit  670 . The output of dual-control multiplexor  804  is coupled to bit storage unit  688 . 
     In addition in FIG. 11, select and enable logic additionally includes NOR gate  814  and inverter  812 . NOR gate  814  receives a “cut column” (CC) signal on one input and a “cut row” (CR) signal on a second input. CC and CR are also sometimes referred to herein as a column mode select signal and a row mode select signal, respectively. The output of NOR gate  814  carries signal {overscore (C)} while the output of inverter  812  carries signal C. Signals C and {overscore (C)} act as “freeze” indication signals as will be later discussed. {overscore (C)} is coupled to switch  816  in multiplexor  802  and switch  820  in multiplexor  804 . Two additional signals are provided: TCLK LO  and TCLK HI . TCLK LO  is coupled to switch  818  in multiplexor  802  and TCLK HI  is coupled to switch  822  in multiplexor  804 . TQ, an output from bit storage unit  688 , is a test data output. 
     As shown, FIG. 11 represents only the functional operation of the test circuitry. When the circuit is implemented at the transistor level and optimized, however, actual separate implementation of all logic elements shown in FIG. 11 may not be necessary. For instance, FIG. 12 shows one specific implementation of STLB  444  with test circuitry as shown in FIG. 11, where the circuitry in FIG. 12 has been optimized. As can be seen by a comparison to FIG. 7, only slight modifications to the circuit of FIG. 7 need be made in order to implement test circuitry in each STLB  444 . Specifically, only three additional transmission gates  1002 ,  1004 , and  1006  have been added in FIG. 12 to obtain the functionality shown in FIG.  11 . Some modifications have also been made to select and enable logic. Specific details of FIG. 12 will not be further discussed as they will be clear to one of skill in the art upon a comparison of FIG. 12 with FIG.  11 . It should further be clear to one of ordinary skill in the art that, although logic gates and inverters are shown, the logic gates in FIG.  12  and other figures are implemented with various transistor configurations. Moreover, one skilled in the art will recognize in FIG.  12  and other figures that various additions, deletions and/or rearrangements of inverters and/or logic gates will still result in an equivalent circuit. 
     Referring again to FIG. 11, in operation, when {overscore (C)} is a logical high (C is a logical low), switches  816  and  820  in multiplexors  802  and  804  are selected, and the circuit behaves in a normal fashion, performing whatever combinational, sequential, or memory function it has been configured to perform as determined by the user-designed circuit. When {overscore (C)} is a logical low signal (C is a logical high), the STLB  444 , is in a freeze mode of operation, freezing (or capturing) the state of the circuit and preventing the bits stored in the STLB from responding to user-designed circuit inputs, and behaves as a master-slave flip-flop. When in a freeze mode of operation ({overscore (C)} is a logical low), the test clock, TCLK, distributed through the signals TCLK LO  and TCLK HI , shifts test data from the data input on line TD through bit storage units  670  and  688  and out through TQ. In one embodiment of the invention TCLK LO  and TCLK HI  are inverse signals to one another, being derived from the same clock (i.e., when TCLK LO  is high, TCLK HI  is low and vice versa), and both TCLK LO  and TCLK HI  only distribute TCLK when the array is in a freeze mode of operation, otherwise leaving switches  818  and  822  open. 
     Essentially then and referring to device  902  in FIG. 13, the STLB  444  will behave somewhat like the device  304  (FIG. 3) used in clock-based scan. Like the device of FIG. 3, the device of FIG. 14 also switches between two interfaces. Rather than simply switching between two interfaces for a single flip-flop, however, when the interfaces to the device  902  are switched, the circuitry internal to the device  902  is reconfigured, although the same circuitry is used. As discussed previously, the function of STLB  444  can be varied simply by varying its inputs. Thus, when FREEZE is selected, the circuitry in STLB  444  that implements a user-designed function (e.g., combinational, sequential, or memory) in normal mode, represented by the “?” in the lower portion  904  of the device  902 , is reconfigured to implement a master-slave flip-flop, represented in the upper portion  906  of device  902 , having test data in (TD), a Test Clock (TCLK) (the clock from which TCLK LO  and TCLK HI  are derived) and test data out (TQ). When FREEZE is selected by placing a logical high signal on line C (FIG.  11 ), STLB  444  becomes non-responsive to regular STLB input signals (e.g., DS, DA), so that except for responding to the test clock (TCLK LO  and TCLK HI ) and test data input signal (TD) the two internal bit storage units do not change state. 
     Addressable Mode Control 
     In an array  410  of function blocks  420 , the STLB  444  of each computation module  440  will be selectable for a normal mode of operation or for a freeze mode of operation by selecting the appropriate column and row mode select signals for the STLB  444 . In other words, in the array  410 , the STLBs  444  form a sub-array where each STLB  444  is addressable by row and column. The column (CC) and row (CR) signals together produce signal C, determining if the STLB is in freeze mode or normal mode. If the column and row mode select signals for the module are both a logical low signal, the STLB will operate in normal mode, performing whatever function (e.g., combinational, sequential, memory) the STLB  444  is configured to perform by the user-designed circuit. If either of the column or row mode select signals is a logical high, then the STLB is in freeze mode. Thus by deasserting both column and row mode select lines, the behavior of any one STLB  444  can be selected for testing in normal mode. 
     For instance, FIG. 14 shows a portion of the sub-array  1210  formed by the STLBs  444   ij  of array  410  (stage one logic blocks  442  and communication modules  460  are not shown in FIG. 14) illustrating the STLBs as flip-flops (i.e., if referring to FIG. 13, the upper interface  906  for each STLB is illustrated). NOR gate  814  and inverter  812  are shown in FIG. 14 simply as OR gate  814 ′ having one input coupled to a column mode select line  1212   i , and a second input coupled to a row mode select line  1216   j . By placing a logical low on each of Column  1  Mode Select line  1212   1  and Row  1  Mode Select line  1216   1 , and by placing logical high values on each of Column  0  Mode Select line  1212   0  and Row  0  Mode Select line  1216   0 , only STIB  444   11  is selected to operate in normal mode. STLBs  444   00 ,  444   01 , and  444   10  will be frozen, retaining as stimulus data that data that was shifted in while all STLBs were frozen. STLB  444   11  will compute the result of whatever stimulus data it receives, e.g., from STLBs  444   00 ,  444   01 , and  444   10  and then capture the result when it is frozen by lines  1212   1  and  1216   1  being asserted. Thus, any STLB  444   ij  can be selectively addressed and tested for proper operation in normal mode. This ability to select any one STLB for exercise during testing of a circuit implemented by the array  410  is sometimes referred to herein as “addressable mode control.” 
     Testing 
     In order to control the state of the circuit implemented by an array  410 , stimulus values are placed in all STLBs in the array  410 . To do so, the flip-flops in each column  1220   i  of STLB sub-array  1210  are “daisy-chained” together. Referring to FIG. 14, in Column  1   1220   1 , stimulus values are first input into flip-flop  1101   10  in STLB  444   10 . The Q output of flip-flop  1011   10  is coupled to the D input of flip-flop  1101   11 . The daisy-chain continues to the bottom of the column  1220   1 . Likewise, the flip-flops of STLBs in Column  0   1220   0  (e.g., STLBs  444   00  and  444   01 ) would be similarly daisy-chained. 
     To place the circuit into a known state, all rows and columns are first selected to freeze by applying logical high values to all of the row freeze lines  1216   j  and/or all of the column freeze lines  1212   i . Stimulus values are placed in a shift register  1502 , where the output of each stage  1504   i  of the shift register  1502  is coupled to the TD input of the first flip-flop  1101   i0  in each column, in one embodiment of the invention. Applying a test clock (TCLK)  1534 , test stimulus values are applied one at a time, from shift register stages  1504   i  to the respective daisy-chains where they are simultaneously shifted through each column  1220   i  via the daisy-chains. A selected STLB  444   ij  would then be selected for normal mode operation by placing a logical low value on both of the row and column freeze lines corresponding to the selected STLB ij . 
     Likewise, to observe the result of a circuit under test, the array of flip-flops, and particularly those in the STLB having just been selected for normal mode, would be frozen via the appropriate row and/or column mode select lines and the resulting data shifted out, again by applying TCLK  1534 . The resulting values are shifted into shift register  1522 , in one embodiment of the invention, from which the values are shifted out of the IC and observed. Other embodiments of the invention may shift data into or out of one column at a time rather than all columns simultaneously and/or may not require use of shift registers  1502 ,  1512 . In one embodiment, as shown, when shift registers are included, the shift registers allow for both serial and parallel test data access to the array. 
     Partition Test 
     Generally, when testing a circuit with one test method in accordance with the invention, STLBs  444  that are connected together in the user-designed circuit should not be tested simultaneously. As discussed above, each STLB can be selectively addressed to operate in normal mode while other STLBs simultaneously remain in freeze mode. Still, testing one STLB block  444  at a time is inefficient and time consuming, particularly when there may be thousands of STLB blocks  444  on a single IC. Testing each STLB block individually would entail repeatedly freezing all STLBs, shifting in test stimulus values, exercising the one STLB selected (i.e., placing the selected STLB in normal mode by de-asserting C), and shifting out all the resulting values from the STLBs. 
     To avoid the above scenario, in one method of testing a gate array in accordance with the invention, herein referred to as “partition test,” the circuit is partitioned to test simultaneously as many STLBs  444  as possible that are not “adjacent”. STLBs  444  are considered to be “adjacent” if the output from one will influence the state of the other. For instance, two STLBs are “adjacent” if (1) there are connection lines directly from the Q output of one STLB to any input of a second STLB (except SX, because SX will not cause any change in values stored in the bit storage units), or (2) there is a connection such as in (1) but with the additional interposition of combinational logic from stage one logic  442 , inverter stage logic  464 , driver stage logic  462 , or a combinational SX-to-Q path through an STLB. 
     By way of example, FIG. 23 shows six STLBs  2444   1-6  connected in a user-designed circuit as indicated by solid lines. The dashed lines indicate which STLBs are “adjacent”. STLB  2444   1  is adjacent to STLB  2444   4  as there is only an inverter between the two STLBs. STLB  2444   2  is adjacent to STLB  2444   4  and STLB  2444   5  as there is only driver logic and combinational logic between the STLBs, respectively. Likewise, STLB  2444   3  is adjacent to STLB  2444   5 . STLB  2444   4  is directly connected to STLB  2444   6  and is thus adjacent. Finally, STLB  2444   5  is adjacent to STLB  2444   6  as it is connected via the SX input of STLB  2444   4 , causing only combinational logic to reside between the two. 
     A first partition is determined that includes the maximum set of STLBs  444  that are not adjacent, such as STLBs  2444   1 ,  2444   2 ,  2444   3 , and  2444   6  in FIG.  23 . After shifting in appropriate stimulus values as described above, all the STLBs  444  in the partition are exercised (placed in normal mode) while the rest of the STLBs  444  in the array are left frozen to maintain the stimulus data. Once the STLBs of the first partition are tested (including exercising and shifting out the resulting captured values), then a next partition is tested, such as  2444   4  and  2444   5 . The partitions are determined by graph coloring techniques that are generally known in the art. 
     More specifically with reference to partition test, FIG. 15 shows a slightly larger portion of the sub-array  1210  of STLBs  444 , with column mode select lines  1212   i  and row mode select lines  1216   j , where there is an STLB at each intersection. The logic for each STLB is not shown for clarity of illustration. As shown, only those STLBs shown with an “X” are part of an exemplary first partition and are to be exercised, as these modules will not influence one another during the testing process (they are “nonadjacent” to one another in the user&#39;s design (not shown)). To test those STLBs  444  that are part of the partition shown in FIG. 15, first, all of the STLBs in array  1210  are frozen, e.g., logical highs are applied to all row mode select lines  1216   j  and/or all column mode select lines  1212   i , to cause the STLBs to function as daisy-chained flip-flops. Next, stimulus data is shifted into all of the STLBs in a columnar daisy-chain fashion as described with respect to FIG. 14 by application of TCLK, and herein referred to as a “shift operation.” A shift operation can take place, in accordance with one embodiment of the invention, only after the STLBs have been frozen. 
     Once the shift operation has completed, a partition mask is applied. A partition mask will cause only the selected STLBs (marked with an “X”) to exercise normally, while other STLBs will remain frozen, driving out stimulus values. The partition mask is applied one row at a time. To do so, mode selection values are applied to the column mode selection lines  1212   i  so that logical low values appear only on the columns in the first row where an STLB is to be exercised. Mode selection values are also applied to the row mode selection lines  1216   j  so that only one row, the first row, has a logical low value. So, in the example of FIG. 15, Row  0  is to be tested first. The values applied to Columns  7 : 0  would be 11101111 and the values applied to Rows  7 : 0  would be 11111110, in order to exercise only the STLB in Row  0  at Column  4 . Once the selected STLBs in the Row  0  have been exercised, the selected STLBs in Row  1  will be exercised in a similar manner, e.g., by applying a mask of 10111110 to Col  7 : 0  and applying a mask of 11111101 to Row  7 : 0 . Note that while only one Row is selected to exercise at a time, more than one column may be selected in various embodiments of the invention. As will be recognized by those of skill in the art, other embodiments of the invention can select more than one row at a time while selecting only one column at a time. The cycle repeats for each row until all selected STLBs in all rows have been exercised. Once all STLBs in all rows have been exercised for the partition, all STLBs are again frozen. Then, a shift operation is performed, by applying TCLK and shifting out the data captured in columnar daisy-chain fashion for observation. A new partition is then selected and the procedure is repeated: stimulus data is shifted in, selected STLBs are exercised row by row, and captured data is shifted out. Several partitions may be required so that each STLB is exercised at least once. The inventors of the present invention have found that for a gate array having approximately 40,000 function blocks designed in accordance with the invention, approximately a dozen partitions will be required on average to fully test array  410 . 
     Scan Emulation Test 
     Using an embodiment of the invention as described with respect to FIG. 14, the array of STLBs can also be used to emulate traditional scan testing, herein referred to as “scan emulation test”. To do so, first stimulus data would be shifted into all STLBs as discussed above for partition test, i.e., by applying 1&#39;s to all column mode select lines and/or all row mode select lines to freeze all STLBs and then applying TCLK to shift data in. Next, rather than selecting a partition, all STLBs are simultaneously exercised, i.e., all column mode select lines and all row mode select lines are set to 0. Thus, in scan emulation testing, all STLBs are either frozen to accept stimulus data (or to shift out resulting data) or all STLBs are in normal mode. 
     In this manner, all flip-flops that are used in the circuit will be used for testing, just as in traditional scan techniques. When a system in accordance with the invention is used for scan emulation testing, it will appear to the user to be somewhat like clock-based scan. However, when an STLB is configured, according to the user-designed circuit, to implement in normal mode something other than a sequential function, e.g., a combinational function, an extra bit in addition to that expected for traditional scan will occur in the shifted-in and shifted-out data since all STLBs, regardless of the function they implement in normal mode, will operate as flip-flops while frozen during scan emulation testing in accordance with the invention. Referring to FIG. 16, the daisy-chained test flip-flops  1610 ,  1612 , and  1614  may be in STLBs that in normal mode operate as a flip-flop  1616 , an AND gate  1618 , and a flip-flop  1620 . In traditional scan, a tester would not expect to be able to place a bit in the STLB containing AND gate  1618  (combinational logic). Once recognized, an extra bit can be inserted into test vectors and/or the extra bit appearing in results can be ignored to enable evaluation in accordance with scan techniques. 
     Enhanced Shift Operation 
     As will be recognized by those of skill in the art, in conventional techniques, including scan techniques, the flip-flops in the user&#39;s design frequently cannot be easily tested without adding additional clock steering circuitry since flip-flops are edge triggered devices and test stimulus data can only generate static patterns. For instance, referring again to FIG. 10, the master latch  740  loads (is transparent to) data incoming on its D input  732  while the clock input  734  is low. After the rising edge of the clock signal, data is captured in the slave latch  750 , which loads when the clock signal is high. Because the flip-flop only responds to clock edges, the internal operation of the flip-flop  730  cannot be easily tested with traditional techniques without additional test structures. For instance, when the clock signal is a static logical low, whatever value is on the D input  732  to the master latch  740  is loaded, but nothing will load into the slave latch  750 . Since only the Q output  736  of the slave latch  750  is observable in traditional scan techniques, the effect of any changes solely to the master latch  740  are not observable. When the clock signal is a static logical high, the master latch  740  will not load—only the slave latch  750  will load the master latch  740  output. Unfortunately, shifting the test data has already loaded the slave from the master so the slave load will have no observable result. Any defects manifesting as faults internal to each flip-flop, therefore, will not be easily isolated with traditional techniques unless care is taken to maintain adequate external controllability of clock signals in order to introduce clock edge transitions from outside the circuit. 
     Nonetheless, using a system clocked by a single TCLK in accordance with the invention, the master and slave latches can each contain independent values during shift operation. In other words, the flip-flop latches can be fully controlled and observed using only logical low and logical high levels, without requiring clock edges. Moreover, when the STLB is configured to perform a memory function in a normal mode of operation, the two latches store independent memory bits, and therefore it is desirable that they be independently controlled and observed during test. 
     FIG. 17 shows use of the STLBs  444   ij  of FIG. 14 redrawn to show the master latch  1102  and the slave latch  1104  of the flip-flop  1101 . Master latch  1102  can generally be identified with bit storage unit  670  (FIG. 11) and slave latch  1104  can generally be identified with bit storage unit  688  (FIG.  11 ). 
     To shift independent values into each master and slave latch, a system in accordance with the invention separately gates TCLK to each master and slave latch. TCLK LO  and TCLK HI  are the signals derived from TCLK once gated. More details will be described with respect to FIG.  18 . 
     Referring now to FIG. 18, a more detailed block diagram of a system in accordance with the invention is described. Similar to that shown in FIG. 14, a 2×2 array of STLBs ( 444   ij ) is shown in FIG. 18 drawn to show the master and slave latch of each flip-flop. The flip-flops, and thus the latches, are daisy-chained together as described with respect to FIG.  14 . Each STLB  444   ij  has an OR gate  814 ′ ij  coupled to a respective row mode select line  1216   j  and a column mode select line  1212   i . The output “C” of each respective OR gate,  814 ′ ij  is coupled to the FREEZE inputs of each of the master latch  1102   ij  and slave latch  1104   ij  for the respective STLB. 
     In addition, various test support logic is shown. Such logic would appear in one embodiment of the invention in periphery region  430  of ASIC  400  (FIG.  4 ). Test support logic includes shift register  1502 , shift register  1512 , shift register  1522 , and test controller  1570 . 
     Shift register  1502  is composed of flip-flops  1504   i . Each stage of the shift register, i.e., each flip-flop  1504   i , is coupled to the data input of the respective master latch  1102   ij  in the first row (Row  0 ) of the respective column. For example, flip-flop  1504   0  is coupled to the TD input of master latch  1102   00 . Likewise, flip-flop  1504   1  is coupled to the test data input of the master latch  1102   10 . 
     Each flip-flop  1504   i  in shift register  1502  is further coupled to the input of an OR gate  1506   i . The second input to each OR gate  1506   i  is coupled to CC 1 . The output of each OR gate  1506   i  is coupled to the first input of an AND gate  1508   i . The second input to each AND gate  1508   i  is coupled to the CCO line. The output of each AND gate  1508   i  is the column mode select signal (CC)  1212   i . 
     A shift register  1522 , including flip-flops  1524   i , is further included. Each flip-flop  1524   i  is coupled to receive the TQ output of the slave latch  1104   ij  in the last row of each column. 
     Test support logic for the rows also includes a shift register and combinational logic similar to that provided for the columns. More specifically, a shift register  1512  is composed of master-slave flip-flops  1514   j . Each stage of the shift register, i.e., each flip-flop  1514   j , is coupled to the input of an OR gate  1516   j . The second input to each OR gate  1516   j  is coupled to CR 1   1546 . The output of each OR gate  1516   j  is coupled to the first input of an AND gate  1518   j . The second input to each AND gate  1518   j  is coupled to CR 0   1544 . The output of each AND gate  1518   j  is the row mode select signal (CR)  1216   j . 
     Clock control logic is further provided. Clock control logic includes, for each row, an AND gate  1526   j , AND gate  1528   j , and an inverter  1530   j . Each AND gate  1526   j  has a first input coupled to a SHIFT signal line  1532  and a second input coupled to inverter  1530   j . The input of inverter  1530   j  is coupled to a TCLK signal line  1534 . The output of each AND gate  1526   j  produces TCLK LO  and is coupled to the G input of each master latch  1102   ij  in its respective row. Each AND gate  1528   j  is coupled to DELAYED SHIFT signal line  1533 . The second input to each AND gate  1528   j  directly to the TCLK signal line  1534 . The output of AND gate  1528   j  provides TCLK HI  and is coupled to each slave latch  1104   ij  in its respective row. 
     To obtain the DELAYED SHIFT signal on line  1533 , SHIFT  1532  is coupled to the D input of latch  1535 , while the gate of latch  1535  is coupled to TCLK  1534 . The Q-output of the latch  1535  is coupled to the DELAYED SHIFT signal line  1533 . In one embodiment, this latch  1535  would be part of a test controller  1570  that controls all of the test signals in FIG. 18 (e.g., CC 0 , CC 1 , CR 0 , CR 1 , TCLK, SHIFT, and DELAYED SHIFT). 
     Each AND gate  1526   j ,  1528   j  further includes an enable input (which may actually be a third input in a 3-input AND gate). The enable input to each AND gate  1526   j  is coupled to the Q-output of each slave latch  1538   j  (i.e., the Q-output of each flip-flop  1514   j ). The enable input to each AND gate  1528   j  is coupled to the Q-output of each master latch  1536   j . 
     Unlike in traditional testing methods where a single value is captured by the master-slave flip-flop, by utilizing clock control logic, independent values can be shifted into and out of each master latch  1102   ij  and slave latch  1104   ij , thereby operating in an “enhanced shift operation.” For instance, when shifting test stimulus values into each column, the shift register  1502  is used to provide the respective test data. Once the data is shifted through the column into the appropriate latch, the clock to that row of latches is turned off. For example, data values can be shifted into the latches until the appropriate data value reaches a chosen slave latch  1104   ij . Then TCLK HI  to that slave latch is turned off. On the next shift through the latches, an independent data value can then be shifted into the master latch for the same STLB. Then TCLK LO  to that master latch can be turned off. Other latches in the column, whose TCLK LO  or TCLK HI  signals have not been turned off, will continue to shift. In this manner, independent values can be placed in each of the master and slave latches for all STLBs in the array. 
     More specifically and by way of example, in FIG. 18 during a shift operation, shift register  1512 , also clocked with TCLK, is first loaded with all logical high values, enabling TCLK to be distributed to all TCLK LO  and TCLK HI  signals in the array. When a desired value for slave latch  1104   01  has been shifted through Column  0  so that it has been shifted into slave latch  1104   01 , then the TCLK HI    1542   1  for Row  1  is turned off by shifting a logical low value into the master latch  1536   1 , disabling AND gate  1528   1 . On the next shift of stimulus values through Column  0 , an independent value will be shifted into master latch  1102   01  and the slave latch  1538   1  will receive a logical low, disabling AND gate  1526   1  and thereby turning off TCLK LO    1540   1  to the master latches  1104   i1  in Row  1 . Logical low values are shifted progressively through the master and slave latches of shift register  1512 , disabling the TCLK LO  and TCLK HI  signals one by one for each row until all the signals  1540   j  and  1542   j  to each of the master and slave latches  1102 ,  1104  have been turned off. 
     To observe the data in both the master and slave latches  1102   ij ,  1104   ij  independently, a similar mechanism is used to shift data out of the array. First, shift register  1512  is loaded with logical low values, preventing the distribution of TCLK to TCLK LO  and TCLK HI  in all of the rows. Then, logical high values are progressively loaded into the shift register  1512 . For instance, the first logical high value will be loaded into master latch  1536   1 , enabling AND gate  1528   1  and turning on TCLK HI  in Row  1 . The data stored in the slave latches  1104   i1  in Row  1  will be shifted out to register  1522 . A logical high value is then received in slave latch  1538   1 , enabling AND gate  1526   1 , and thus TCLK LO  in Row  1 . Data stored in the master latches  1102   i1  in Row  1  will be shifted out (via the Row  1  slave latches) and received by shift register  1522 . Each TCLK HI  and TCLK LO  is sequentially enabled. At the time all of them have been enabled, half of the data will have been shifted out of the array. The remaining half of the data is shifted out with all enabled, i.e., with TCLK gated onto each row. 
     In FIG. 18, CR 0   1544 , CR 1   1546 , CC 0   1548 , and CC 1   1550  are mode lines that determine the operating mode of the array. For instance, when all of the mode lines are a logical low, the circuit operates in normal mode (no STLB is frozen). If CC 0  and CC 1  are both logical high&#39;s (and/or if CR 0  and CR 1  are both logical high&#39;s), the array is in freeze mode and can, by assertion of TCLK, perform a shift operation or an enhanced shift operation—shifting test stimulus values into or out of the array. If CC 0  is a logical high, CC 1  is a logical low, CR 0  is a logical high, and CR 1  is a logical low, the STLBs are enabled for addressable mode control: values shifted into the shift registers  1502  and  1512  will control which STLBs are to be in freeze mode and which are to operate in normal mode. 
     Note that shift registers  1502  and  1512  serve multiple purposes. Shift register  1502  is used to shift in stimulus values in a shift operation, but, once stimulus values are shifted into the STLBs, then the shift register can be used for other purposes, e.g., for addressable mode control. Likewise, shift register  1512  is used for clock control during shift operation, but subsequent to the completion of shift operation, shift register  1512  can also be used for addressable mode control. 
     As described with reference to FIG. 5, other logic may form part of any circuit formed from array  410 . For instance various combinational circuits (e.g., stage one logic blocks  442  or inverter stages  464 ) may lie between any STLBs  444 . Nonetheless, the inventors have found that when using and designing circuits in accordance with their invention, typically only 3-4 non-STLB stages will be linked together and that ¼ to ⅓ of all logic used is mapped into the STLBs of a function block  420  in accordance with the invention. Thus, using a system in accordance with the invention for implementing test circuitry and testing a gate array will easily achieve good fault coverage without user design modifications. 
     As should be understood by those of skill in the art, while an array is formed by function blocks  420 , an array is also formed by STLBs  444 . The term “sub-array” as used herein is only used for clarity of description and should not be construed to limit the invention. As should also be understood, an array in accordance with an embodiment of the invention could be formed completely with logic blocks formed in accordance with the principles described with respect to STLBs and without other interspersed logic such as stage one logic, inverter stages or driver stages. 
     Tri-state Contention Prevention 
     Nevertheless, if driver stages  462  are included, an embodiment of a system in accordance with the invention further prevents signals driven by such driver stages  462  from fighting with one another during a shift operation. FIG. 19 shows an embodiment of the driver stage  462  of communication module  460 . Tri-state buffer  1602  receives signal DI as a first input and produces signal Z. Tri-state buffer  1602  further receives signal EI to its enable input. 
     FIG. 20 is a more detailed drawing showing a specific implementation of an embodiment of driver stage  462  shown in FIG.  19 . Tri-state buffer  1602  is implemented with p-channel transistor  1704 , n-channel transistor  1706 , NAND gate  1708 , NOR gate  1710 , and inverters  1712  and  1714 . Signal EI is input to inverter  1714  whose output is coupled to one input of NAND gate  1708  and to inverter  1712 . The output of inverter  1712  is coupled to one input of NOR gate  1710 . Input signal DI is coupled to the second input of NAND gate  1708  and the second input of NOR gate  1710 . The output of NAND gate  1708  is coupled to the gate of transistor  1704 . The output of NOR gate  1710  is coupled to the gate of transistor  1706 . The drain of transistor  1704  is coupled to the drain of transistor  1706 , forming output Z. 
     FIG. 21 shows two tri-state drivers  1602   A  and  1602   B , such as that shown in FIG. 19, where the outputs of the drivers are coupled together on line  1802  and where each tri-state driver has each of its inputs coupled to a respective flip-flop  1801   j  (i.e., from an STLB) when the STLBs are frozen. Although not shown for purposes of clarity of the figure, flip-flops  1801   j  are coupled in a daisy-chain manner when the STLBs are frozen (in normal mode, and as has been discussed throughout the detailed description, the respective STLBs may perform a function other than the sequential function shown). Suppose that it is desirable to test the ability of tri-state driver  1602   B  to drive out a logical low value on line  1802 . To do so, the pattern (X 100 ) shown within the flip-flops  1801   1-4  would need to be shifted into the flip-flops, where X is a “don&#39;t care” value. The value A (either a logical high or a logical low) will be shifted into flip-flop  1801   5 , the next flip-flop in the column. However, on the clock shift immediately prior to the final clock shift, which shifts in the described pattern, the flip-flops will hold the pattern  100 A: both tri-state drivers  1602   A  and  1602   B  will be enabled and they will cause contention on line  1802  if A≠1, which will be true 50% of the time. 
     In accordance with an embodiment of the invention, to prevent contention, during a shift operation, all tri-state outputs are held to a logical high value. FIG. 22 shows an additional input {overscore (TH)} added to the circuit of FIG.  20 . Inverter  1712  (FIG. 20) is replaced with NAND gate  1712 ′ and receives as one input {overscore (TH)} and as a second input, the output from inverter  1714 . An additional p-channel transistor  1804  is coupled in parallel to transistor  1704 . The gate of transistor  1804  is also coupled to input {overscore (TH)}. When {overscore (TH)} is a logical high, the circuit will function as in FIG.  20 . When {overscore (TH)} is brought to a logical low, a logical high (V DD ) signal will be driven on Z. By coupling {overscore (TH)} to all tri-state drivers and by bringing {overscore (TH)} to a logical low when shift operation is to be entered, contention on tri-state outputs will be prevented during shift operation. 
     A system in accordance with the invention has thus been described that has several advantages over traditional techniques used for testing ASICs. First, the test circuitry in each STLB is predesigned within each function block and is thus not dependent on the ultimate user-designed circuit implemented by the gate array nor does the designer need to insert test circuitry himself. 
     The logic of particular function blocks also need not be dedicated solely to testing, but can be used to perform both test and normal operating functions—in other words, the STLB circuitry is reused for freeze and normal modes of operation. STLBs do not even need to implement sequential logic in the user-designed circuit to still be able to control and observe data in each STLB—i.e., even combinational logic is testable without having to insert extra flip-flops for testing purposes. Yet, only minimal additional transistors need to be added to the function blocks  420  to implement testing in accordance with the invention. 
     Further, each STLB is equipped with addressable mode control, allowing each STLB in each function block to be individually exercised. Nonetheless, the entire circuit implemented by the gate array can be tested relatively quickly using the dynamic partitioning of partition test. 
     A gate array designed in accordance with the invention can also emulate traditional scan techniques, allowing those circuit designers more comfortable with traditional scan testing to test the gate array in that manner. 
     Finally, the test clock signal can be independently enabled and disabled to each of the master and slave latches that form flip-flops used for test purposes, permitting the ability to independently shift test data into and out of the master and slave latches of a test flip-flop. Using latches and a TCLK in accordance with the invention also eliminates most DFT rules conventionally used in designing ICs since problems associated with testing edge-triggered flip-flops are eliminated (all testing is easily implemented with static patterns). 
     Although headings have been used in this description, they are to serve as a guide to the reader only and should not be construed to limit the invention. 
     It should be understood that the particular embodiments described above are only illustrative of the principles of the present invention, and various modifications could be made by those skilled in the art without departing from the scope and spirit of the invention. Thus, the scope of the present invention is limited only by the claims that follow.