Patent Publication Number: US-2013234638-A1

Title: Power converter for driving switched reluctance motor

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims benefit under 35 U.S.C. 119 of JP2012-048906 filed on Mar. 6, 2012, the title of TRANSVERSE FLUX MACHINE APPARATUS, JP2012-85172 filed on Apr. 4, 2012, the title of POWER CONVERTER FOR DRIVING SWITCHED RELUCTANCE MOTOR, JP2012-90645 filed on Apr. 12, 2012, the title of POWER CONVERTER FOR DRIVING SWITCHED RELUCTANCE MOTOR and JP2012-95387 filed on Apr. 19, 2012, the title of POWER CONVERTER FOR DRIVING SWITCHED RELUCTANCE MOTOR, the entire content of which is incorporated herein reference. 
     BACKGROUND OF INVENTION 
     1. Field of the Invention 
     The present invention relates to a power converter for driving a switched reluctance motor, in particular a power converter for driving a switched reluctance motor having phase windings of four or six or more than six of even number. 
     2. Description of the Related Art 
     It is known for a switched reluctance motor (SRM) to have many advantages for a variable-speed application such as a traction motor, in particular a direct-drive hub motor. However, it is known for the SRM to have drawbacks such as acoustic noise, vibration, torque ripples and a torque/weight ratio in comparison with a popular permanent magnet synchronous motor. It is known that acoustic noise, vibration, torque ripples of the SRM is reduced by increasing phase number of the SRM. However, it is not easy for the power converter to increase the phase number because of a cost of the multi-phase power converter. Many circuit topologies of power converters are proposed for SRMs. 
     U.S. Pat. No. 7,906,931 describes a power converter with a full-bridge per phase. However, The full-bridge power converter requires four switches per phase. A popular asymmetric bridge converter shown in  FIG. 1  requires two switches per phase. Three phase windings  3 U,  3 V and  3 W are driven independently by the power converter having an upper bridge  9 A and a lower bridge  9 B. The upper bridge  9 A has three legs consisting of upper switches T 1 , T 3  and T 5  and lower diodes D 1 , D 3  and D 5 . The lower bridge  9 B has three legs consisting of lower switches T 2 , T 4  and T 6  and upper diodes D 2 , D 4  and D 6 . 
       FIG. 2  shows a Miller converter having less transistors. The Miller converter has a common leg  9   c  consisting of a switch T 8  and a diode D 8  instead of the lower bridge  9 B shown in  FIG. 1 . However, the Miller converter cannot execute demagnetization of one phase and magnetization of another phase simultaneously.  FIG. 3  shows an arranged Miller converter having a lower switch T 8 B instead of the diode D 8  shown in  FIG. 2 .  FIG. 4  shows a four-phase capacitor split power converter, which is one kind of a power converter using two voltage sources. The capacitor split converter has four switches T 1 -T 4 , four diodes and two capacitors C 1  and C 2 . A neutral point N is connected to an upper DC link line  1000  via a pair of a X-phase winding  3 X and an upper switch T 1  or a pair of a Z-phase winding  3 Z and an upper switch T 3  or a capacitor C 1 . Further, the neutral point N is connected to a lower DC link line  2000  via a pair of a Y-phase winding  3 Y and a lower switch T 2  or a pair of a T-phase winding  3 T or a lower switch T 4  and a capacitor C 2 . Capacitors C 1  and C 2  keep a voltage of neutral point N to a half of a DC link voltage. 
     However, the voltage split four phase power converter shown in  FIG. 4  needs voltage split capacitors C 1  and C 2  having a large capacity and a high cost due to apply a half of the DC link voltage to each phase windings  3 X- 3 T. 
     A energy-absorber of a power converter is known. The energy-absorber has a capacitor for accumulating a demagnetizing current temporally. Typically, the energy-absorber connects each of upper diodes D 2 , D 4  and D 6  to a DC link. However, it is difficult to employ the energy absorber for the power converter of driving a large SRM because the capacitor having a large capacity under of high voltage type is large and expensive. 
     CITATION LIST 
     Patent Literature 
     
         
         PTL 1: U.S. Pat. No. 7,906,931 
       
    
     SUMMARY OF INVENTION 
     An object of the invention is to provide a simple SRM drive capable of reducing acoustic noise, vibration and torque ripples of a multi-phase SRM. Another object of the invention is to provide a simple SRM drive capable of extending a speed range of a multi-phase SRM. Another object of the invention is to provide a simple SRM drive capable of increasing a torque/weight ratio. Another object of the invention is to provide a control method of a multi-phase SRM drive having benefits mentioned above. 
     As for the invention, a power converter having an upper bridge and a lower bridge, which are controlled by a controller, drives a switched reluctance machine having phase windings of four or six or more than six of even number. Each upper leg of the upper bridge is connected to each upper phase windings connected to an upper neutral point. Each lower leg of the lower bridge is connected to each lower phase windings connected to an lower neutral point. The upper leg has a pair of a lower switch and an upper diode connected in series. The lower leg has a pair of an upper switch and a lower diode connected in series. The upper neutral point is connected to the lower neutral point directly or via at least one of a connection switch and a connection diode. 
     The power converter further has a current-adjusting circuit having at least one transistor for adjusting a neutral current flowing from the upper neutral point to the lower neutral point. Therefore, it is capable of constructing the SRM with low acoustic noise, low vibration and low torque ripples without using an expensive power converter having many power transistors or a large voltage-split capacitors. It is known that acoustic noise, vibration and torque ripples are reduced by means of increasing phase number. The power converter having essentially one switch per phase is capable of driving a six-phase SRM because the transistor of the current-adjusting circuit adjusts phase currents. 
     According to a preferred embodiment, the upper bridge magnetize three of odd numbered stator poles to a first magnetic polarity, and the lower phase windings magnetize three of even numbered stator poles a second magnetic polarity. Therefore, an iron loss of the six-phase SRM is reduced because the SRM of radial flux type has short flux passages 
     According to another preferred embodiment, one phase current of one bridge with three legs is equal to a sum of two phase currents of another bridge with three legs in an asymmetric mode. Therefore, the simple power converter can drive the six-phase SRM. 
     According to another preferred embodiment, one bridge supplies an increasing current of one phase and a decreasing current of another phase. The other bridge supplies an essentially constant current of another phase in the asymmetric mode. Therefore, the simple power converter without voltage split capacitors can drive the six-phase SRM. 
     According to another preferred embodiment, the two bridges supply each phase current having an essentially trapezoid waveforms. Therefore, the current difference between the two bridges is reduced. 
     According to another preferred embodiment, the two bridges supply each phase current exciting each magnetic flux having essentially half rectified sinusoidal waveforms to each phase winding. For example, the two bridges supply each phase current having essentially half rectified sinusoidal waveforms. Or, the two bridges apply to each phase voltage having essentially half rectified sinusoidal waveforms to each phase winding. Therefore, the current difference between the two bridges is reduced. Moreover, acoustic noise, vibration are reduced. Further, an iron loss of the six-phase SRM is reduced. Similarly, other known SRMs, for example a three-phase SRM, can have a low iron loss by means of employing each phase magnetic flux having essentially half rectified sinusoidal waveforms. The reason that each phase currents having half rectified sinusoidal waveforms reduce the iron loss is explained hereinafter. A predetermined average value of phase current must be supplied for one magnetization period of a SRM in order to produce a predetermined average value of a motor torque. First, the phase current is increased from zero to a predetermined value, and the phase current is decreased from the predetermined value to zero. A hysterics loss is similar to a friction loss on the mechanics. The hysterics loss is increased, when a changing speed of magnetic flux and a magnetic flux density are high. The changing speed of the magnetic flux with half rectified sinusoidal waveforms is lower than the other waveforms, when the magnetic flux density is high. Therefore, the iron loss of a SRM is reduced, when the phase currents having the half rectified sinusoidal waveforms are supplied to the SRM. In other words, changing of the magnetic flux density is easy, when the magnetic flux density is low, but the changing of the magnetic flux density is difficult, when the magnetic flux density is high. Preferably, the phase currents with half rectified sinusoidal waveforms is supplied to a SRM, when a rotation speed of the SRM is high because the iron loss of a variable-speed SRM such as the traction motor is increased very much in the high speed area. It is capable of applying the phase voltages having essentially half rectified sinusoidal waveforms to phase windings of a SRM instead of supplying the phase currents having essentially half rectified sinusoidal waveforms. 
     According to another preferred embodiment, the current-adjusting circuit has a current-absorbing leg connected to the upper neutral point and a current-supplying leg connected to the lower neutral point. Therefore, voltage ripples of the neutral points are reduced even though a current difference between two bridges becomes large. Therefore, the current difference between the two bridges is compensated without the voltage split capacitors. 
     According to another preferred embodiment, the current-absorbing leg has a current-absorbing switch for absorbing the current from the upper neutral point. The current-supplying leg has a current-supplying switch for supplying the current to the lower neutral point. Therefore, the current difference between the two bridges is compensated without the voltage split capacitors. 
     According to another preferred embodiment, the current-absorbing switch and the current-supplying switch are switched in accordance with either of the voltage of the neutral points or a current difference between the upper bridge and the lower bridge in the accelerated bridge mode having an essentially equal voltage of the neutral points. The switches are switched in order to reduce the ripples of the voltage of the neutral points. Therefore, the current difference between the two bridges is reduced. 
     According to another preferred embodiment, the upper bridge and the current-absorbing leg constitutes one Miller converter in a dual Miller mode when the connection switch is turned off. Similarly, the lower bridge and the current-supplying leg constitutes the other Miller converter in the dual Miller mode. Therefore, a torque is increased in the dual Miller mode, because a full voltage of the DC power source is applied to two Miller converters each. 
     According to another preferred embodiment, the dual Miller mode is selected, when either of the two bridges has a trouble. Therefore, the reliability of the power converter is improved. According to another preferred embodiment, the dual Miller mode is selected, when a rotation speed of the SRM is a high speed area. Therefore, the SRM produces a sufficient torque in the high speed area even though the back electromagnetic force (EMF) is increased in the high speed area. 
     According to another preferred embodiment, the magnetizing mode and the demagnetizing mode of one bridge are executed alternately with a predetermined frequency in the dual Miller mode. Therefore, the magnetization speed and the demagnetization speed are improved. 
     According to another preferred embodiment, changing between the magnetizing mode and the demagnetizing mode is executed by means of switching the current-supplying switch and the current-absorbing switch. Therefore, the magnetization speed and the demagnetization speed are improved. 
     According to another preferred embodiment, each of phase current consists of a DC current component and a sinusoidal AC current component. An amplitude of the DC current component is essentially equal to an amplitude of the sinusoidal AC current component. Therefore, an iron loss is reduced in a high speed area. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a circuit topology configuration showing a prior three-phase asymmetric bridge converter. 
         FIG. 2  is a circuit topology configuration showing a prior three-phase Miller converter. 
         FIG. 3  is a circuit topology configuration showing an prior arranged three-phase Miller converter with a common half-bridge. 
         FIG. 4  is a circuit topology configuration showing a prior four-phase capacitor split power converter. 
         FIG. 5  is a circuit topology configuration showing a six-phase power converter of a first embodiment for driving a six-phase SRM. 
         FIG. 6  is a schematic side view showing a stator with twelve stator poles. 
         FIG. 7  is a schematic development showing a six-phase radial flux SRM of 12/14 type shown in  FIG. 6 . 
         FIG. 8   FIG. 8  is a timing chart showing waveforms of phase currents and inductances of the six-phase SRM shown in  FIGS. 6 and 7 . 
         FIG. 9  is a flow chart showing an operation of the power converter shown in  FIG. 5 . 
         FIG. 10  is a schematic side view showing a short flux passages in the SRM shown in  FIG. 6 . 
         FIG. 11  is a schematic side view showing the short flux passages in the SRM shown in  FIG. 6 . 
         FIG. 12  is a schematic axial cross-section showing a six-phase transverse flux switched reluctance machine having tandem structure. 
         FIG. 13  is a circumferential development showing arrangement of stator teeth of the six-phase TFSRM shown in  FIG. 12 . 
         FIG. 14  is a circumferential development showing arrangement of rotor teeth of the six-phase TFSRM shown in  FIG. 12 . 
         FIG. 15  is a timing chart showing waveforms of phase currents and inductances of an arranged six-phase SRM shown in  FIG. 16 . 
         FIG. 16  is a schematic development showing a six-phase radial flux SRM of 12/10 type. 
         FIG. 17  is a timing chart showing half rectified sinusoidal waveforms of another six-phase current for driving the six-phase SRM 
         FIG. 18  is a circuit topology configuration showing a six-phase power converter of a second embodiment for driving a six-phase SRM. 
         FIG. 19  is a timing chart showing waveforms of phase currents of the power converter driven with an accelerated bridge mode. 
         FIG. 20  is a circuit topology configuration showing a magnetization mode of a dual Miller mode in a first sub period. 
         FIG. 21  is a circuit topology configuration showing a demagnetization mode of the dual Miller mode in the first sub period. 
         FIG. 22  is a circuit topology configuration showing another magnetization mode of the dual Miller mode in a second sub period. 
         FIG. 23  is a circuit topology configuration showing another demagnetization mode of the dual Miller mode in the second sub period. 
         FIG. 24  is a timing chart showing a switching pattern of the power converter driven with the asymmetric bridge mode. 
         FIG. 25  is a timing chart showing a switching pattern of the power converter driven with the accelerated bridge mode. 
         FIG. 26  is a timing chart showing a switching pattern for the power converter driven with the dual Miller mode. 
         FIG. 27  is a flow chart showing one control example of power converter shown in  FIG. 18 . 
         FIG. 28  is a timing chart showing six phase currents having waveforms being equal each to a sum of a DC current and an AC current with a sinusoidal waveforms. 
         FIG. 29  is a flow chart for selecting a silent mode supplying the phase currents shown in  FIG. 28 . 
         FIG. 30  is a circuit topology configuration showing an arrangement of the power converter shown in  FIG. 18 . 
         FIG. 31  is a circuit topology configuration showing another arrangement of the power converter shown in  FIG. 18 . 
         FIG. 32  is a circuit topology configuration showing another arrangement of the power converter shown in  FIG. 18 . 
         FIG. 33  is a circuit topology configuration showing another arrangement of the power converter shown in  FIG. 18 . 
         FIG. 34  is a timing chart showing waveforms of phase currents and inductances of a three-phase SRM having six phase windings shown in  FIG. 35 . 
         FIG. 35  is a schematic development showing a three-phase SRM of 6/4 type having six phase windings connected to a neutral point each. 
         FIG. 36  is a schematic development showing a three-phase SRM of 6/8 type having six phase windings connected to a neutral point each. 
         FIG. 37  is a timing chart showing three-phase current supplied to the three-phase SRM having six phase windings shown  FIG. 36 . 
         FIG. 38  is a schematic axial cross-section showing a three-phase transverse tandem flux switched reluctance machine having two phase windings wound of the same phase. 
         FIG. 39  is a circumferential development showing arrangement of stator teeth of the three-phase TFSRM shown in  FIG. 38 . 
         FIG. 40  is a circumferential development showing arrangement of rotor teeth of the three-phase TFSRM shown in  FIG. 38 . 
         FIG. 41  is a circuit topology configuration showing a four-phase power converter of a third embodiment for driving a four-phase SRM. 
         FIG. 42  is a circuit topology configuration showing another four-phase power converter for driving the four-phase SRM. 
         FIG. 43  is a schematic timing chart showing inductances and currents in the asymmetric bridge mode of the four-phase power converter shown in  FIGS. 41-42 . 
         FIG. 44  is a schematic timing chart showing inductances and currents in the dual bridge mode of the four-phase power converter shown in  FIGS. 41-42 . 
         FIG. 45  is a circuit topology configuration showing a current-adjusting circuit operated in the asymmetric mode of the four-phase power converter shown in  FIGS. 41-42 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A First Embodiment 
     The first embodiment is explained referring to  FIGS. 6-21 .  FIG. 6  is a circuit topology configuration showing a six-phase power converter  9  for driving a six-phase SRM having six phase windings  3 U 1 - 3 W 2  shown in  FIGS. 7 and 8 .  FIG. 7  is a schematic cross-section showing one example of the six-phase SRM of 12/14 type.  FIG. 8  is a schematic development showing a stator  2  having six stator poles  20  and a rotor  4  having seven rotor poles  40 . The SRM shown in  FIG. 7  has two sets of six phase windings  3 U 1 - 3 W 2  wound respectively on the six stator poles  20  in turn. The six phase windings  3 U 1 - 3 W 2  consist of the U 1 -phase windings  3 U 1 , the U 2 -phase winding  3 U 2 , the V 1 -phase windings  3 V 1 , the V 2 -phase winding  3 V 2 , the W 1 -phase windings  3 W 1  and the W 2 -phase winding  3 W 2 . 
     The power converter  9  consists of an upper bridge  9 A, a lower bridge  9 B and a controller  300 . The upper bridge  9 A has a U 1 -phase leg  901 , a V 1 -phase leg  903  and a W 1 -phase leg  905 . The U 1 -phase leg  901  consists of an upper switch T 1  and a lower diode D 1  connected in series. The V 1 -phase leg  903  consists of an upper switch T 3  and a lower diode D 3  connected in series. The W 1 -phase leg  905  consists of an upper switch T 5  and a lower diode D 5  connected in series. 
     Upper ends of the upper switches T 1 , T 3  and T 5  are connected to a high potential DC link line  1000 . Lower ends of the lower diodes D 1 , D 3  and D 5  are connected to a low potential DC link line  2000 . A connection point of U 1 -phase leg  901  is connected to one end of the U 1 -phase winding  3 U 1 . A connection point of V 1 -phase leg  903  is connected to one end of the V 1 -phase winding  3 V 1 . A connection point of W 1 -phase leg  905  is connected to one end of the W 1 -phase winding  3 W 1 . The other ends of the phase windings  3 U 1 ,  3 V 1  and  3 W 1  are connected to an upper neutral point NU. 
     The lower bridge  9 B has a U 2 -phase leg  902 , a V 2 -phase leg  904  and a W 2 -phase leg  906 . The U 2 -phase leg  902  consists of a lower switch T 2  and an upper diode D 2  connected in series. The V 2 -phase leg  904  consists of an lower switch T 4  and an upper diode D 4  connected in series. The W 2 -phase leg  906  consists of a lower switch T 6  and an upper diode D 6  connected in series. Lower ends of the lower switches T 2 , T 4  and T 6  are connected to a low potential DC link line  2000 . Upper ends of the upper diodes D 2 , D 4  and D 6  are connected to the high potential DC link line  1000 . A connection point of U 2 -phase leg  902  is connected to one end of the U 2 -phase winding  3 U 2 . A connection point of V 2 -phase leg  904  is connected to one end of the V 2 -phase winding  3 V 2 . A connection point of W 2 -phase leg  906  is connected to one end of the W 2 -phase winding  3 W 2 . The other ends of the windings  3 U 2 ,  3 V 2  and  3 W 2  are connected to a lower neutral point NL. 
     A star-connected upper three-phase winding  3   k  consists of three phase windings  3 U 1 ,  3 V 1  and  3 W 1 . A star-connected lower three-phase winding  3 L consists of three phase windings  3 U 2 ,  3 V 2  and  3 W 2 . As shown in  FIG. 6 , three phase windings  3 U 1 ,  3 V 1  and  3 W 1  of the upper three-phase winding  3 K are wound on odd numbered stator poles  20  respectively. Similarly, three phase windings  3 U 2 ,  3 V 2  and  3 W 2  of the lower three-phase winding  3 L are wound on even numbered stator poles  20  respectively. Upper three-phase windings  3   k  magnetize odd numbered stator pole  20  to N-poles. Lower three-phase windings  3 L magnetize even numbered stator pole  20  to S-poles. 
     A motor-driving method of power converter  9  is explained referring to  FIG. 8 .  FIG. 8  is a timing chart showing six phase currents IU 1 -IW 2  supplied to phase windings  3 U 1 - 3 W 2 . In  FIG. 8 , each of the phase windings  3 U 1 - 3 W 2  has the largest inductance values LL of inductances LU 1 -LW 2 . At each point ‘a’ shown in  FIG. 8 , inductances LU 1 -LW 2  have the smallest inductance value each. 
     The U 1 -phase current IU 1  flows from the leg  901  to U 1 -phase winding  3 U 1 . The V 1 -phase current IV 1  flows from the leg  903  to V 1 -phase winding  3 V 1 . The W 1 -phase current IW 1  flows from the leg  905  to W 1 -phase winding  3 W 1 . The U 2 -phase current IU 2  flows from U 2 -phase winding  3 U 2  to the leg  902 . The V 2 -phase current IV 2  flows from V 2 -phase winding  3 V 2  to the leg  904 . The W 2 -phase current IW 2  flows from W 2 -phase winding  3 W 2  to the leg  906 . 
     In  FIG. 8 , each phase has a magnetization period, a demagnetizing period and an absent period executed in turn. Each magnetization period starts at the time point ‘a’ and finishes at a time point ‘g’. Each demagnetization period starts at the time point ‘g’ and finishes at a time point ‘n’. Each magnetization period consists of a current-increasing period and a constant current period. Each of the current-increasing periods starts at time point ‘a’ and finishes at time points ‘e, p, q, r’. Each of the constant current periods starts at time points ‘e, p, q, r’ and finishes at time point ‘g’. 
     Each real line passing on time points ‘a, r, m and n’ shows each phase current having a current amplitude I 1 . Each real line passing on time points ‘a, q, k and n’ shows each phase current having a current amplitude I 2 . Each real line passing on time points ‘a, p, j and n’ shows each phase current having a current amplitude I 3 . Each real line passing on time points ‘a, e, i and n’ shows each phase current having a current amplitude I 4 . Each real line passing on time points ‘a, b, d, e, f, h, i and n’ shows each phase current having a current amplitude I 5 . Each real line passing on time points ‘a, b, c, d, e, f, g, h, i and n’ shows each phase current having a current amplitude I 6 . 
     Each of phase currents IU 1 -IW 2  increases in each current-increasing period. However, phase currents IU 1 -IW 2  with the amplitudes I 5  and I 6  have a part of the constant current period in the current-increasing period from the time point ‘a’ to the time point ‘e’. Each of phase currents IU 1 -IW 2  is mostly constant in each constant current period from the points ‘e, p, q, and r’ to the points ‘i, j, k, and m’. However, phase currents IU 1 -IW 2  with the amplitudes I 5  and  16  are not constant in the constant current period from the point ‘e’ to the time point ‘i’. Each of phase currents IU 1 -IW 2  decreases in each current-decreasing period from the points ‘i, j, k, and m’ to the point ‘n’. 
     Each current-increasing periods has sixty degrees of electric angle. Each of constant-current periods has sixty degrees of electric angle. Each of current-decreasing periods has sixty degrees of electric angle. Each phase difference between adjacent two phase currents has sixty degrees of electric angle. It is important that a sum of an increasing phase current in the current-increasing period and a decreasing phase current in the current-decreasing current is equal to a constant current in the constant current period. 
     Phase currents IU 1 -IW 2  is supplied by means of PWM-switching the switches T 1 -T 6 . In a first case, switches T 1 -T 6  are PWM-switched in the current-increasing periods and the constant current periods. In a second case, switches T 1 -T 6  are PWM-switched in the current-increasing periods and the current-decreasing periods. It should be considered that the real lines  11 - 16  shown in  FIG. 8  show schematic current configurations. Further, the real lines passing on the time point ‘i’ shows the largest demagnetizing current passing through diodes D 1 -D 6 . 
     Therefore, phase current IW 2  becomes equal to a sum of phase currents IU 1  and IW 1  in a sub period ‘A’ from a time point t 4  to a time point t 5 . Phase current IU 1  becomes equal to a sum of phase currents IW 2  and IU 2  in a sub period ‘B’ from a time point t 5  to a time point t 6 . Phase current IU 2  becomes equal to a sum of phase currents IU 1  and IV 1  in a sub period ‘C’ from a time point t 6  to a time point t 1 . Phase current IV 1  becomes equal to a sum of phase currents IU 2  and IV 2  in a sub period ‘D’ from a time point t 1  to a time point t 2 . Phase current IV 2  becomes equal to a sum of phase currents IV 1  and IW 1  in a sub period ‘E’ from a time point t 2  to a time point t 3 . Phase current IW 1  becomes equal to a sum of phase currents IV 2  and IW 2  in a sub period ‘F’ from a time point t 3  to a time point t 4 . 
     After all, it is considered that a voltage of the neutral points NU and NL becomes a half of DC link voltage continuously, when two of six switches T 1 -T 6  are PWM-switched in order to accord the phase current in the constant-current period to a sum of the adjacent two phase currents in the current-increasing period and the current-decreasing period. As the result, the large capacitors C 1  and C 2  shown in  FIG. 4  are abbreviated or becomes very small. Moreover, power converter  9  having only six switches T 1 -T 6  can drives a six-phase SRM with low acoustic noise and low vibration. 
     Accordingly, the simple power converter  9  reduces the acoustic noise, the vibration and the torque ripples largely because magnetic force between stator  2  and rotor  4  are dispersed spatially and sequentially. Further, power converter  9  supplies three phase currents simultaneously by means of PWM-switching only two of switches T 1 -T 6 . Therefore, the switching power loss of power converter  9  is reduced. Furthermore, power losses of diodes D 1 -D 6  are reduced because the demagnetizing current flows through only one diode. In prior arts shown in  FIG. 1-2 , the demagnetizing current flows through two diodes in turn. 
       FIG. 9  is a flow chart showing the switch control. First, information for controlling the SRM is detected at a step S 100 . The information includes an torque instruction value Tin, a rotor position Protor, a rotor speed Vrotor and phase currents IU 1 -IW 2 . At next step S 102 , inductances LU 1 -LW 2  is searched from a memorized map showing a relation between the phase inductance LU 1 -LW 2 , phase currents IU 1 -IW 2  and the rotor position Protor. At next step S 104 , phase torques TU 1 -TW 2  are calculated in accordance with phase currents IU 1 -IW 2 , phase inductances LU 1 -LW 2  and the rotor speed Vrotor. Then, a total torque Ttotal is calculated in accordance with six phase torques TU 1 -TW 2 . 
     Further, a torque difference Tdif between the total torque Ttotal and the torque instruction value Tin is calculated. At next step S 106 , next values of phase currents IU 1 -IW 2  are decided in accordance with torque difference Tdif and the detected phase currents IU 1 -IW 2 . At next step S 108 , gate voltages of switches T 1 -T 9  are decided in accordance with next phase currents IU 1 -IW 2 . Instead of the above soft feedback operation, a hard feedback operation can be adopted. 
       FIG. 10  is a schematic side view showing magnetic flux Fx of the six-phase SRM in the sub period B having sixty degrees of electric angle.  FIG. 12  is a schematic side view showing the magnetic flux Fx in the sub period C having sixty degrees of electric angle. The magnetized odd numbered stator poles  20  have N-poles. The magnetized even numbered stator poles  20  have S-poles. Magnetic flux Fx circulate via adjacent only two stator poles  20  by employing the three phase currents supplied simultaneously. In other words, the six-phase SRM shown in  FIGS. 10 and 11  becomes so-called the short flux path SRM without employing special core structure. Therefore, an iron loss is reduced largely. Instead of a conventional six-phase SRM of 12/14 type, another known six-phase is SRM, for example a six-phase SRM with U-shaped segmented rotor cores or a six-phase SRM with U-shaped segmented stator cores can be employed. 
     A First Arranged Embodiment 
     The first arranged embodiment is explained referring to  FIGS. 12-14 .  FIG. 12  is a schematic cross-section showing another six-phase transverse flux switched reluctance machine (TFSRM) having six single-phase TFSRMs arranged in tandem to an axial direction AX. Each of the six single-phase TFSRMs has each of rotor cores  4 U 1 - 4 W 2  facing to each of stator cores  2 U 1 - 2 W 2 . Each of the rotor cores  4 U 1 - 4 W 2  has the left teeth  40 L and the right teeth  40 R connected with a ring-shaped back core. Each of the stator cores  2 U 1 - 2 W 2  has the left teeth  20 L and the right teeth  20 R connected with a ring-shaped back core. Each of ring-shaped phase windings  3 U 1 - 3 W 2  is accommodated in each ring-shaped slot formed between each pair of the left stator teeth  20 L and the right stator teeth  20 R. The left rotor teeth  40 L face the left stator teeth  20 L in the radial direction RA. The right rotor teeth  40 R face the right stator teeth  20 R in the radial direction RA. 
       FIG. 13  is a circumferential development showing arrangement of stator teeth  20 L and  20 R.  FIG. 14  is a circumferential development showing arrangement of rotor teeth  40 L and  40 R. The left stator teeth  20 L, the right stator teeth  20 R, the left rotor teeth  40 L and the right rotor teeth  40 R are arranged to the circumferential direction PH each. The power converter  9  shown in  FIG. 5  can drive the six-phase TFSRM shown in  FIGS. 12-14 . Configurations of currents IU 1 -IW 2  and inductances LU 1 -LW 2  are shown in  FIG. 8 . 
     A Second Arranged Embodiment 
     The second arranged embodiment is explained referring to  FIGS. 15 and 16 .  FIG. 15  is a timing chart showing another six-phase SRM shown in  FIG. 16 . The six-phase 12/10 SRM shown in  FIG. 16  has twelve stator poles per ten rotor poles. Six phase currents IU 1 -IW 2  shown in  FIG. 15  are same as six phase currents IU 1 -IW 2  shown in  FIG. 8 . However, inductances LU 1 -LW 2  shown in  FIG. 16  have different configurations from inductances LU 1 -LW 2  shown in  FIG. 8  because a number of rotor poles  40  are different to each other. 
     A Third Arranged Embodiment 
     The third arranged embodiment is explained referring to  FIG. 17 .  FIG. 17  is a timing chart showing another example of six phase currents IU 1 -IW 2 . Real lines show phase currents IU 1 -IW 2  with small amplitude. Broken lines show phase currents IU 1 -IW 2  with large amplitude. Six inductances LU 1 -LW 2  show the inductances of phase windings  3 U 1 - 3 W 2  of six-phase SRM shown in  FIGS. 6-14 . Six inductances LU 1 ′-LW 2 ′ show the inductances of phase windings  3 U 1 - 3 W 2  of six-phase SRM shown in  FIGS. 15-16 . Each of phase currents IU 1 -IW 2  has a positive half of sinusoidal waveform each. U 1 -phase current IU 1  only flows from time point t 4  to time point t 1 . U 2 -phase current IU 2  flows from time point t 5  to time point t 2 . V 1 -phase current IV 1  flows from time point t 6  to time point t 3 . V 2 -phase current IV 2  flows from time point t 1  to time point t 4 . W 1 -phase current IW 1  flows from time point t 2  to time point t 5 . W 2 -phase current IW 2  flows from time point t 3  to time point t 6 . 
     In sub period D, phase current IW 2  is equal to a sum of phase currents IU 1  and IW 1 . In sub period E, phase current IU 1  is equal to a sum of phase currents IW 2  and IU 2 . In sub period F, phase current IU 2  is equal to a sum of phase currents IU 1  and IV 1 . In sub period A, phase current IV 1  is equal to a sum of phase currents IW 2  and IV 2 . In sub period B, phase current IV 2  is equal to a sum of phase currents IV 1  and IW 1 . In sub period C, phase current IW 1  is equal to a sum of phase currents IV 2  and IW 2 . Therefore, vibration and acoustic noise are reduced. The configurations of phase current IU 1 -IW 2  are formed by means of PWM-switching of two phases. 
     Furthermore, an iron loss is reduced largely by means of employing the phase currents IU 1 -IW 2  having the half rectified sinusoidal waveforms each. In the prior SRM-driving method, it is unknown to drive a switched reluctance motor (SRM) with phase currents having the half rectified sinusoidal waveforms each. Further, it is unknown to reduce the iron loss by means of driving the SRM with phase currents having the half rectified sinusoidal waveforms each. It is desirable to supply the phase currents with the half rectified sinusoidal waveforms to a SRM rotating in a high speed area because the iron loss is reduced largely in the high speed area. Similarly, an iron loss of the other known SRM is reduced by means of employing the phase currents with the half rectified sinusoidal waveforms. It is capable of applying phase voltages with the half rectified sinusoidal waveforms to phase windings of a SRM in order to supply phase currents having essentially half rectified sinusoidal waveforms to the phase windings of the SRM. Moreover, it is capable of modulating each phase current in accordance with non-linear magnetic characteristic of the magnet core in order to excite each phase magnetic flux having essentially half rectified sinusoidal waveforms. 
     A Second Embodiment 
     The second embodiment is explained referring to  FIGS. 18-32 .  FIG. 18  is a circuit topology configuration showing another six-phase power converter  9 . The power converter  9  shown in  FIG. 18  is essentially same as power converter  9  shown in  FIG. 5  except a neutral voltage controller  9 C shown in  FIG. 18 . The neutral voltage controller  9 C shown in  FIG. 18  consists of a connection switch T 9 , a current-absorbing leg  907  and a current-supplying leg  908 . 
     The connection switch T 9  connects the upper neutral point NU of the upper three-phase winding  3 K to the lower neutral point NL of the lower three-phase winding  3 L. The current-absorbing leg  907  has a current-absorbing diode D 7  and a current-absorbing switch T 7  connected in series. A cathode electrode of the current-absorbing diode D 7  is connected to the high potential DC link line  1000 . An anode electrode of current-absorbing diode D 7  is connected to upper neutral point NU. The current-absorbing switch T 7  connects the upper neutral point NU to the low potential DC link line  2000 . 
     The current-supplying leg  908  has a current-supplying switch T 8  and a current-supplying diode D 8  connected in series. The current-supplying switch T 8  connects lower neutral point NL to the high potential DC link line  1000 . An anode electrode of the current-supplying diode D 8  is connected to low potential DC link line  2000 . An cathode electrode of current-supplying diode D 8  is connected to lower neutral point NL. 
     The controller  300  controls motor-driving operation of power converter  9 . Controller  300  has three motor-driving modes, which are called an asymmetric bridge mode, an accelerated bridge mode and a dual Miller mode. The asymmetric bridge mode is executed, when the connection switch T 9  is turned on, and the switch T 7  and T 8  are turned off. Therefore, the asymmetric bridge mode is same as the motor operation explained referring to  FIGS. 5-17 . 
     (The Accelerated Bridge Mode) 
     The accelerated bridge mode is explained referring to  FIG. 19 . In the accelerated bridge mode, the connection switch T 9  is turned on. Further, either of the current-absorbing switch T 7  and the current-supplying switch T 8  is turned-on in order to reduce a difference between the constant current Ic and the sum of the increasing current Ii and the decreasing current Id. In other words, either of the switches T 7  and T 8  is turned-on in order to reduce a difference between a second-phase current and a sum of a first-phase current and a third-phase current. The difference between the second-phase current and the sum of the first-phase current and the third-phase current is equal to either of a current I 7  of the switch T 7  and a current I 8  of the switch T 8 . Either of the switches T 7  and T 8  is PWM-switched in order to accord either of the currents I 7  and I 8  with the difference between the second-phase current and the sum of the first-phase current and the third-phase current. Therefore, power converter  9  is capable of supplying phase currents IU 1 -IW 2  with a large amplitude in order to produce a large torque. 
     According to the accelerated bridge mode, the current difference between upper bridge  9 A and lower bridge  9 B is absorbed by means of PWM-switching either of current-absorbing switch T 7  and current-supplying switch T 8 . Therefore, the current difference between upper bridge  9 A and lower bridge  9 B is absorbed by either of currents I 7  and I 8 . 
       FIG. 19  is a timing chart showing three phase currents IU 1 , IU 2  and IW 2  in sub periods D-F. Real lines show three phase currents IU 1 , IU 2  and IW 2  with large amplitudes. Broken lines show three phase currents IU 1 , IU 2  and IW 2  with small amplitudes. For example, the increasing currents Ii of phase currents IU 1 , IU 2  and IW 2  are supplied with the so-called one-pulse method or the single-pulse method. In the one-pulse method, a one-pulse of the gate voltage is applied to a gate electrode of switches. 
     The PWM-switching method can be employed instead of the one pulse method. It is considered that the difference between U 2 -phase current IU 2  and the sum of U 1 -phase current IU 1  and V 1 -phase current IV 1  becomes zero by means of PWM-switching of the increasing current of V 1 -phase current IV 1  in sub period F. However, the current difference Ix between U 2 -phase current (the second phase current) IU 2  and the sum of U 1 -phase current (the first phase current) IU 1  and V 1 -phase current (the third phase current) IV 1  has large ripples in the large current operation. The controller  300  calculates the current difference Ix in accordance with the memorized map and the detected information, and supplies the current difference Ix to phase windings  3 U 1 - 3 W 2  by means of PWM-switching current-absorbing T 7  and current-supplying switch T 8 . In  FIG. 19 , the switch T 7  is PWM-switched in periods T 7 . The switch T 8  is PWM-switched in periods T 8 . At time points Tx, the current difference Ix becomes zero. 
     A feedback control method can be employed in order to control the switches T 7  and T 8 . Switch T 7  is turned on, when a sum of phase currents IU 1 , IV 1  and IW 1  is larger than a sum of phase currents IU 2 , IV 2  and IW 2 . Similarly, switch T 8  is turned on, when the sum of phase currents IU 1 , IV 1  and IW 1  is smaller than the sum of phase currents IU 2 , IV 2  and IW 2 . 
     According to a preferred embodiment executing the accelerated bridge mode, the switches T 7  and T 8  can be switched with the feed back control method in accordance with a neutral voltage of neutral points NU and NL in order to keep the neutral voltage to a half of the DC link voltage. The switch T 7  is turned on when the neutral voltage becomes higher than a half of the DC link voltage. Similarly, the switch T 8  is turned on when the neutral voltage becomes lower than the half of the DC link voltage. 
     (The Dual Miller Mode) 
     The dual Miller mode of the motor-driving operation is explained referring to  FIGS. 20-23 . According to the dual Miller mode, connection switch T 9  is turned off. In other words, a pair of upper bridge  9 A and current-absorbing leg  907  constitutes a first Miller converter. Another pair of lower bridge  9 B and current-supplying leg  908  constitutes a second Miller converter. 
     The fundamental motor operation of the Miller converter is explained again referring to  FIG. 2 . In the magnetizing period, the magnetizing current of one phase flows by means of the turning-on of the switch T 8  and one of switches T 2 , T 4 , T 6 . In the magnetizing period, a freewheeling current of another phase flows through the switch T 8  and another of diodes D 2 , D 4  and D 6 , when the demagnetization of another phase in not completed yet. In the demagnetizing period, the demagnetizing current of one phase flows by means of turning-off all switches T 2 , T 4 , T 6  and T 8 . The magnetizing period of one phase and the demagnetization period of another phase cannot be overlapped to each other. Thus, the conventional Miller mode has a drawback that the demagnetization of the freewheeling current is slow in a low speed area because the back EMF is small. 
       FIGS. 20 and 21  show phase currents IU 1 , IW 1  and IW 2  in the sub period D shown in  FIG. 9 .  FIG. 20  shows the magnetization mode executed in sub period D. The switches T 1 , T 7 , T 6  and T 8  are turned on. The magnetizing current IU 1  flows through U 1 -phase winding  3 U 1  via the switches T 1  and T 7 . The freewheeling current IWI circulates through W 1 -phase winding  3 W 1  via diode D 5  and switch T 7 . The constant current IW 2  flows through W 2 -phase winding  3 W 2  via the switches T 6  and T 8 . 
       FIG. 21  shows the demagnetization mode executed in sub period D. The switches T 8  and T 6  are turned on, and the switch T 7  is turned off. The demagnetizing current IW 1  charges the DC power source (not shown) via DC link lines  1000  and  2000 . Decreasing of the freewheeling current IU 1  is slow. The constant current IW 2  flows through W 2 -phase winding  3 W 2  via the switches T 6  and T 8 . The magnetization modes and the demagnetization modes in sub periods F and B are essentially same as the magnetization mode and the demagnetization mode in sub periods D explained above. 
       FIGS. 22 and 23  show phase currents IU 1 , IU 2  and IW 2  in sub period E.  FIG. 22  shows the magnetization mode in sub period E. In  FIG. 22 , the switches T 1 , T 2 , T 7  and T 8  are turned on. The magnetizing current IU 2  flows through U 2 -phase winding  3 U 2  via the switches T 2  and T 8 . The freewheeling current IW 2  circulates through W 2 -phase winding  3 W 2  via diode D 6  and switch T 8 . The constant current IU 1  flows through U 1 -phase winding  3 U 1  via the switches T 1  and T 7 . 
       FIG. 23  shows the demagnetization mode in the sub period E. In  FIG. 23 , the switches T 1  and T 7  are turned on, and the switch T 8  is turned off. The demagnetizing current IW 2  charges the DC power source (not shown) via DC link lines  1000  and  2000 . Decreasing of the freewheeling current IU 2  is slow. The constant current IU 1  flows through U 1 -phase winding  3 U 1  via the switches T 1  and T 7 . The magnetization mode and the demagnetization mode in sub periods A and C are essentially same as the magnetization mode and the demagnetization mode in sub periods E explained above. 
     According to the above dual Miller mode of the second embodiment, the magnetization mode and the demagnetization mode are executed alternately in each sub period with a predetermined frequency. Preferably, either of bridges  9 A and  9 B supplies both of the increasing current (magnetizing current) Ii and the decreasing current (demagnetizing current) Id. The other one of bridges  9 A and  9 B supplies the constant current Ic. For example, both of the increasing current Ii and the constant current Id are supplied with the PWM-switching. Thus, both of the magnetization of one phase and the demagnetization of another phase are executed well in each sub period. It is understand that each of executing times of the magnetization and the demagnetization in the dual Miller mode becomes half in comparison with a asymmetric bridge mode. However, the DC voltage applied to each of phase windings  3 U 1 - 3 W 2  becomes double. Accordingly, both of the magnetization speed and the demagnetization speed are not delayed by means of repeating the magnetization and the demagnetization alternately with a predetermined carrier frequency. 
     For example, the magnetization mode shown in  FIG. 20  and the demagnetization mode shown in  FIG. 21  are repeated alternately in each of sub periods D, F and B by means of PWM-switching the switch T 7 . U 1 -phase current IU 1  becomes the freewheeling current in the magnetization mode because the switch T 7  is turned off instead of turning-off of the switch T 1 . Thus, reduction of the U 1 -phase current IU 1 , which is the magnetizing current, is suppressed. The magnetization current IW 2 , which is the constant current Ic, is kept to a predetermined constant value by means of PWM-switching either or both of the upper switch T 8  and the W 2 -phase lower switch T 6 . 
     Similarly, the magnetization mode shown in  FIG. 22  and the demagnetization mode shown in  FIG. 23  are repeated alternately in each of sub periods E, A and C by means of PWM-switching the switch T 8 . U 2 -phase current IU 2  becomes the freewheeling current in the magnetization mode because the switch T 8  is turned off instead of turning-off of the switch T 2 . Thus, reduction of the U 2 -phase current IU 2 , which is the magnetizing current, is suppressed. The magnetization current IU 1 , which is the constant current Ic, is kept to a predetermined constant value by means of PWM-switching either or both of the lower switch T 7  and the U 1 -phase upper switch T 1 . 
     It is important that the demagnetization current is the largest at each initial time of sub periods A-F, and the magnetization current is the largest at each final time of sub periods A-F. Accordingly, controller  300  decreases a ratio Rt (=a demagnetization time Tde/a magnetization time Tma) continuously during each of sub periods A-F. Therefore, the average value of the magnetizing current and the average value of the demagnetizing current are not reduced by means of the above time-sharing operation. Thus, applying the full battery voltage applied to each phase in the dual Miller mode increases the average values of the magnetizing current and the demagnetizing current. In other words, the demagnetizing modes shown in  FIGS. 21 and 23  are executed longer than the magnetizing modes shown in  FIGS. 20 and 22  in each initial stage of sub periods A-F. The demagnetizing time becomes gradually short, and the magnetizing time becomes longer gradually. 
     Switching patterns of switches T 1 -T 9  in the above three modes are shown in  FIGS. 24-26 .  FIG. 24  is a timing chart showing the switching pattern of the switches T 1 -T 9  in the asymmetric bridge mode shown in  FIG. 8 .  FIG. 25  is a timing chart showing the switching pattern of the switches T 1 -T 9  in the accelerated bridge mode shown in  FIG. 20 . In  FIGS. 24 and 25 , the current-increasing period Ti and the constant current period Tc are executed in turn. In  FIGS. 24 and 25 , connection switch T 9  is turned on. 
     In  FIG. 24 , switches T 1 -T 6  are PWM-switched during the current increasing period Ti. In  FIG. 25 , switches T 1 -T 6  are PWM-switched during the constant current Tc. Further, the switches T 7  and T 8  are PWM-switched alternately.  FIG. 26  is a timing chart showing switching pattern of the switches T 1 -T 9  for executing the dual Miller mode. Power converter  9  produces two three-phase currents in the dual Miller mode. However, a current ripples IS of a total current supplied from the DC power source have the small amplitude and high frequency, because the two three-phase currents has a phase difference to each other. As the result, a smoothing capacitor connected to power converter  9  becomes small. 
     (A Mode-Changing Method) 
     The mode-changing method is explained referring to  FIG. 27 .  FIG. 27  is a flow chart showing one example of the mode-changing method executed by controller  300 . At a first step S 600 , it is judged whether or not upper bridge  9 A is normal. If upper bridge  9 A has a trouble, only lower bridge  9 B is driven as the Miller converter at a step S 602 . In other words, the magnetization current is supplied from the switch T 8  to one of three lower switches T 2 , T 4  and T 6  through the phase windings  3 U 2 ,  3 V 2  and  3 W 2 . 
     At a next step S 604 , it is judged whether or not lower bridge  9 B is normal. If lower bridge  9 B has a trouble, only upper bridge  9 A is driven as the Miller converter at a step S 606 . In other words, the magnetization current is supplied from one of three upper switches T 1 , T 3  and T 5  to the switch T 7  through the phase windings  3 U 1 ,  3 V 1  and  3 W 1 . Next, it is judged whether or not both of upper bridge  9 A and lower bridge  9 B are normal at a step  608 . If both of upper bridge  9 A and lower bridge  9 B have a trouble each, bridges  9 A and  9 B are stopped, and the controller  300  outputs the alarm signal at a step  614 . 
     Next, it is judged whether or not a detected rotor speed Nr is higher than a predetermined high threshold value Nrthh at a step S 610 . When the rotor speed Nr is higher than the high threshold value Nrthh, the dual Miller mode is executed at a step S 612 . The dual Miller mode is excellent for driving the SRM in the high-speed area because a full-scale of battery voltage is applied to each phase winding  3 U 1 - 3 W 2  each. In the high-speed area, phase currents IU 1 -IW 2  of the sufficient value are supplied to phase windings  3 U 1 - 3 W 2  even though the back EMF is increased. 
     Next, it is judged whether or not a detected rotor speed Nr is higher than a predetermined high threshold value Nrthh at a step S 610 . When the rotor speed Nr is higher than the high threshold value Nrthh, the dual Miller mode is selected at a step S 612 . Further, it is judged whether or not a detected motor rotation speed Nr is lower than a predetermined low threshold value NrthL at a step S 616 . When the speed Nr is lower than the low threshold value NrthL, it is judged whether or not an instruction value of the motor torque Ti is larger than a predetermined value Tth as a step S 618 . When the instruction value of the motor torque Ti is not larger than the predetermined value Tth, the asymmetric bridge mode is executed at a step S 620 . When the instruction value of the motor torque Ti is larger than the predetermined value Tth, the accelerated bridge mode is executed at a step S 622 . The asymmetric bridge mode is excellent in the low speed area. The accelerated bridge mode is excellent in the low-speed-high-torque area. 
     A First Arranged Embodiment 
     The first arranged embodiment is explained referring to  FIGS. 28 and 29 .  FIG. 28  is a timing chart showing another configurations of phase currents IU 1 -IW 2  in the dual Miller mode. U 1 -phase current IU 1  is equal to a sum of a DC current Idc and a sinusoidal current IU 1   ac . U 2 -phase current IU 2  is equal to a sum of the DC current Idc and a sinusoidal current IU 2   ac . V 1 -phase current IV 1  is equal to a sum of a DC current Idc and a sinusoidal current IV 1   ac . V 2 -phase current IV 2  is equal to a sum of the DC current Idc and a sinusoidal current IV 2   ac . W 1 -phase current IW 1  is equal to a sum of a DC current Idc and a sinusoidal current IW 1   ac . W 2 -phase current IW 2  is equal to a sum of the DC current Idc and a sinusoidal current IW 2   ac.    
     The configurations of phase currents IU 1 -IW 2  are enable, when the dual-Miller mode is executed, because the sum of the first phase current and the third phase current is not equal to the second phase current. The configurations of phase currents IU 1 -IW 2  shown in  FIG. 28  is desirable for the high-speed area of the SRM because the iron loss of the SRM is reduced largely. The configurations of phase current IU 1 -IW 2  are formed by means of PWM-switching of two phases. 
       FIG. 29  is a flow chart showing selection of the above sinusoidal configurations of phase currents IU 1 -IW 2  shown in  FIG. 28 . First, it is judged whether or not a rotor rotation speed Vr is higher than a predetermined threshold value Vth at a step S 200 . When the rotor speed Vr is not higher, it is judged whether or not a driver hope a silent drive mode at a step S 202 . When the driver hopes a strong torque, the other current configuration is selected at a step S 204 . When the rotor speed Vr is higher or the driver hopes the silent driving, the current configuration shown in  FIG. 28  is employed at a step S 206 . Instead of the current configuration shown in  FIG. 28 , the phase currents having half certificated sinusoidal waveforms shown in  FIG. 16  can be employed in the silent drive mode at the step S 206 . For example, the phase currents of half certificated sinusoidal waveforms shown in  FIG. 16  are employed the in a low speed area or in the asymmetric bridge mode or the accelerated bridge mode, and the phase currents consisting of the sum of DC current and sinusoidal AC current each are employed the in a high speed area or in the dual Miller mode. Therefore, the comfortable driving and high efficiency at the high speed are realized. 
     In the prior SRM-driving method, it is unknown to drive a switched reluctance motor (SRM) with phase currents having a sum of a DC current and a sinusoidal AC current. Further, it is unknown to reduce the iron loss by means of driving the SRM in the high speed area with a sum of a DC current and a sinusoidal AC current. The other SRM, for example the three-phase SRM, can be used the above SRM-driving method employing the sum of the DC current and the sinusoidal AC current. 
     A Second Arranged Embodiment 
     The second arranged embodiment is explained referring to  FIG. 30 . The power converter  9  shown in  FIG. 30  has a connection diode D 9  instead of connection switch T 9 . Power converter  9  shown in  FIG. 30  has mostly same motor-driving operation of as power converter  9  shown in  FIG. 18 . However, a current flows from upper neutral point NU to lower neutral point NL, when a voltage of upper neutral point NU is higher than a sum of a voltage of lower neutral point NL and a voltage drop of diode D 9  in the dual Miller mode of  FIG. 30 . In other words, the currents of current-absorbing switch T 7  and current-supplying switch T 8  are reduced, when the voltage of upper neutral point NU is higher than the sum of the voltage of lower neutral point NL and the voltage drop of diode D 9  in the dual Miller mode of  FIG. 30 . 
     A Third Arranged Embodiment 
     The third arranged embodiment is explained referring to  FIG. 31 . The power converter  9  shown in  FIG. 31  has a connection diode D 9  instead of connection switch T 9 . Further, the current-absorbing diode D 7  and the current-supplying diode D 8  are abbreviated in  FIG. 31 . However, the magnetization is accelerated by means of turning-on the switches T 7  and T 8  because the magnetizing currents flow through the switches T 7  and T 8 . 
     A Fourth Arranged Embodiment 
     The fourth arranged embodiment is explained referring to  FIG. 32 . The switches T 7  and T 8  are abbreviated in  FIG. 32 . However, the demagnetization is accelerated by means of turning-off the connection switch T 9  because the demagnetizing currents flow to the DC power source via the diodes D 7  and D 8 . 
     A Fifth Arranged Embodiment 
     The fifth arranged embodiment is explained referring to  FIG. 33 . In  FIG. 33 , connection switch T 9  and diodes D 7  and D 8  are abbreviated in  FIG. 33 . However, the difference between the current of upper bridge  9 A and the current of lower bridge  9 B in the asymmetric bridge mode is absorbed by means of switching the switches T 7  and T 8 . 
     A Sixth Arranged Embodiment 
     The sixth arranged embodiment is explained referring to  FIGS. 34 and 35 .  FIG. 34  is a timing chart showing phase currents IU 1 -IW 2  supplied to a popular three-phase SRM shown in  FIG. 35 .  FIG. 35  is a schematic development showing the three-phase SRM of 6/4 type. Six phase windings  3 U 1 - 3 W 2  are wound six stator poles  20  respectively and in turn. The rotor core  4  has four rotor poles  40  per six stator poles  20 . U 1 -phase current IU 1  flowing through U 1 -phase winding  3 U 1  is the same as U 2 -phase current IU 2  flowing through U 2 -phase winding  3 U 2 . V 1 -phase current IV 1  flowing through V 1 -phase winding  3 V 1  is the same as V 2 -phase current IV 2  flowing through V 2 -phase winding  3 V 2 . W 1 -phase current IW 1  flowing through W 1 -phase winding  3 W 1  is the same as W 2 -phase current IW 2  flowing through W 2 -phase winding  3 W 2 . 
     The motor operation of the three-phase SRM is essentially same as the motor operation of the second embodiment shown in  FIGS. 18-33 . Accordingly, the three-phase SRM having six phase windings  3 U 1 - 3 W 2  connected to a common neutral point has the same advantages as the six-phase SRM mentioned above. However, U 2 -phase current IU 2  has the same phase as U 1 -phase current IU 1 . V 2 -phase current IV 2  has the same phase as V 1 -phase current IV 1 . W 2 -phase current IW 2  has the same phase as W 1 -phase current IW 1 . In other words, the switches T 1  and T 2  have the same switching pattern. The switches T 3  and T 4  have the same switching pattern. The switches T 5  and T 6  have the same switching pattern. 
     Another difference between the three-phase operation shown in  FIG. 34  and the six-phase operation shown in  FIGS. 5-32  is explained referring to  FIG. 34 . The three-phase SRM of 6/4 type has a first current-decreasing period Td 1  and a second current-decreasing period Td 2  as shown in  FIG. 34 . In the first current-decreasing periods Td 1  of the dual Miller mode, the two Miller converters execute only the demagnetizing operation each. In the second current-decreasing periods Td 2  of the dual Miller mode, the two Miller converters execute the demagnetizing operation and the magnetizing operation alternately. The symmetric bridge mode is preferable in a low speed area. The dual Miller mode is preferable in a high speed area. 
     A Seventh Arranged Embodiment 
     The seventh arranged embodiment is explained referring to  FIGS. 36 and 37 .  FIG. 36  is a schematic development of another three-phase SRM with six stator poles  20  per eight rotor poles  40 . Six phase windings  3 U 1 - 3 W 2  are wound on six stator poles  20  in turn and respectively. It is capable to wind two phase windings with the same phase on one stator pole.  FIG. 37  is a timing chart showing phase currents IU 1 -IW 2  supplied to phase windings  3 U 1 - 3 W 2  of the three-phase SRM shown in  FIG. 36 . The symmetric bridge mode is preferable in a low speed area. The dual Miller mode is preferable in a high speed area. 
     A Eighth Arranged Embodiment 
     The eighth arranged embodiment is explained referring to  FIGS. 38-40 .  FIG. 38  is a schematic cross-section showing a three-phase TFSRM having three single-phase TFSRMs arranged in tandem to an axial direction AX. Each of the three single-phase TFSRMs has each of rotor cores  4 U- 4 W facing to each of stator cores  2 U- 2 W. Each rotor core has the left teeth  40 L and the right teeth  40 R connected with a ring-shaped back core. Each stator core has the left teeth  20 L and the right teeth  20 R connected with a ring-shaped back core. Ring-shaped U-phase windings  3 U 1  and  3 U 2  are accommodated in a ring-shaped slot of U-phase stator cores  2 U. Ring-shaped V-phase windings  3 V 1  and  3 V 2  are accommodated in a ring-shaped slot of V-phase stator cores  2 U. Ring-shaped W-phase windings  3 W 1  and  3 W 2  are accommodated in a ring-shaped slot of W-phase stator cores  2 W. The left rotor teeth  40 L face the left stator teeth  20 L in the radial direction RA. The right rotor teeth  40 R face the right stator teeth  20 R in the radial direction RA. 
       FIG. 39  is a circumferential development showing arrangement of stator teeth  20 L and  20 R.  FIG. 40  is a circumferential development showing arrangement of rotor teeth  40 L and  40 R. The left stator teeth  20 L, the right stator teeth  20 R, the left rotor teeth  40 L and the right rotor teeth  40 R are arranged to the circumferential direction PH each. 
     Power converters  9  explained above is capable of driving the three-phase SRM shown in  FIGS. 38-40 . Configurations of phase currents IU 1 -IW 2  are shown in  FIG. 36 . The symmetric bridge mode is preferable in a low speed area. The dual Miller mode is preferable in a high speed area. 
     A Third Embodiment 
     The third embodiment is explained referring to  FIG. 41 .  FIG. 41  is a circuit topology configuration showing a four-phase power converter  9  for driving a four-phase SRM. The power converter  9  shown in  FIG. 41  is essentially same as power converter  9  shown in  FIG. 18 . However, upper bridge  9 A has only two legs  901  and  903  for driving X-phase winding  3 X and Z-phase winding  3 Z. Similarly, lower bridge  9 B has only two legs  902  and  904  for driving Y-phase winding  3 Y and T-phase winding  3 T. 
     Four-phase power converter  9  shown in  FIG. 41  can have the asymmetric bridge mode, the accelerated bridge mode and the dual Miller mode like the second embodiment explained above. Further, the power converter  9  shown in  FIG. 41  is capable of supplying four-phase currents IX, IY, IZ and IT having the half rectified sinusoidal waveforms each. Moreover, the power converter  9  shown in  FIG. 41  is capable of supplying each phase currents being equal to a sum of a DC current component and a sinusoidal AC current component. Both of amplitudes of the DC current component and the sinusoidal AC current component is equal. Therefore, four-phase power converter  9  shown in  FIG. 41  has similar advantages to six-phase power converter shown in  FIG. 18 . 
     A First Arranged Embodiment 
     The first arranged embodiment is explained referring to  FIG. 42 . Power converter  9  shown in  FIG. 42  is essentially same as power converter  9  shown in  FIG. 41 . However, power converter  9  shown in  FIG. 42  has connection diode D 9  instead of connection switch T 9  shown in  FIG. 41 . The operation and the advantages of power converter  9  shown in  FIG. 42  is mostly same as power converter  9  shown in  FIG. 30 . 
       FIG. 44  is a timing chart showing phase currents IX-IT in the asymmetric mode of the power converter shown in  FIG. 41  or  FIG. 42 . Each of phase currents IX-IT is PWM-switched from each time point tk in each of turning-on periods of the switches T 1 -T 4 . A voltage Vn of the neutral points Nu and NL becomes mostly a half of the DC link voltage Vdclink, when a sum of all currents IX, IY, IZ and IT becomes zero. Accordingly, it is capable of changing the turned-on of the switch T 7  and the turned-on of the switch T 8  at each time point when the sum of all phase currents IX, IY, IZ and IT becomes zero.  FIG. 45  is a timing chart showing phase currents in the dual-Miller mode of the power converter shown in  FIG. 41  or  FIG. 42 .  FIG. 46  is a circuit topology configuration showing an example of changing the turned-on switch T 7  and the turned-on switch T 8  in accordance with the voltage Vn of the neutral points NU, NL in the asymmetric bridge mode shown in  FIG. 43 . A comparator  701  compares the voltage Vn and a reference voltage Vref(=0.5 Vdclink). A gate controller  702  turns on the switch T 7 , when the voltage Vn is higher than the reference voltage Vref. A gate controller  703  turns on the switch T 8 , when the voltage Vn is lower than the reference voltage Vref. 
     Six-phase power converters  9 , which means a power converter having six legs connected to the neutral point, has been explained in the first embodiment and the second embodiment. Four-phase power converters  9  having four legs connected to the neutral point has been explained in the third embodiment. It is easily considered for a skilled engineer that the power converter  9  is capable of having more phases (more legs connected to the neutral point) in order to drive a SRM with more-phases. Further, it is considered easily that power converter  9  is capable of magnetizing a switched reluctance generator (SRG). Furthermore, power converter  9  can include the energy-absorber having a capacitor or a reactor in order to accumulate a residual magnetic energy temporarily. For example, it is capable that anode electrodes of lower diodes D 1 , D 3 , D 5  can be connected to a capacitor capable of absorbing a current from the DC link line  2000  via a reactor or a switch. Similarly, cathode electrodes of upper diodes D 2 , D 4 , D 6  are connected to a capacitor capable of supplying a current to the DC link line  1000  via a reactor or a switch. Further, diodes D 1 -D 9  of power converter  9  can include transistors having essentially same rectification operation. Or, it is capable to connect transistors to diodes D 1 -D 9  in parallel in order to reduce the diode power loss.