Patent Publication Number: US-9425653-B2

Title: Transmitters for wireless power transmission

Description:
CLAIM OF PRIORITY 
     This application is a continuation of U.S. patent application Ser. No. 12/561,069 entitled “TRANSMITTERS FOR WIRELESS POWER TRANSMISSION,” filed Sep. 16, 2009, the disclosure of which is hereby incorporated by reference in its entirety. Application Ser. No. 12/561,069 claimed priority under 35 U.S.C. §119(e) to:
         U.S. Provisional Patent Application 61/098,742 entitled “MAGNETIC POWER USING A CLASS E AMPLIFIER” filed on Sep. 20, 2008, the disclosure of which is hereby incorporated by reference in its entirety.   U.S. Provisional Patent Application 61/097,859 entitled “HIGH EFFICIENCY TECHNIQUES AT HIGH FREQUENCY” filed on Sep. 17, 2008, the disclosure of which is hereby incorporated by reference in its entirety.   U.S. Provisional Patent Application 61/147,081 entitled “WIRELESS POWER ELECTRONIC CIRCUIT” filed on Jan. 24, 2009, the disclosure of which is hereby incorporated by reference in its entirety.   U.S. Provisional Patent Application 61/218,838 entitled “DEVELOPMENT OF HF POWER CONVERSION ELECTRONICS” filed on Jun. 19, 2009, the disclosure of which is hereby incorporated by reference in its entirety.       

    
    
     BACKGROUND 
     1. Field 
     The present invention relates generally to wireless charging, and more specifically to devices, systems, and methods related to portable wireless charging systems. 
     2. Background 
     Typically, each powered device such as a wireless electronic device requires its own wired charger and power source, which is usually an alternating current (AC) power outlet. Such a wired configuration becomes unwieldy when many devices need charging. Approaches are being developed that use over-the-air or wireless power transmission between a transmitter and a receiver coupled to the electronic device to be charged. The receive antenna collects the radiated power and rectifies it into usable power for powering the device or charging the battery of the device. Wireless energy transmission may be based on coupling between a transmit antenna, a receive antenna and a rectifying circuit embedded in the host electronic device to be powered or charged. Transmitters, including transmit antennas, face conflicting design constraints such as relative small volume, high efficiency, a low Bill Of Materials (BOM), and high reliability. Accordingly, there is a need to improve a transmitter design for wireless power transmission which satisfy the various design objectives. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a simplified block diagram of a wireless power transmission system. 
         FIG. 2  illustrates a simplified schematic diagram of a wireless power transmission system. 
         FIG. 3  illustrates a schematic diagram of a loop antenna, in accordance with exemplary embodiments. 
         FIG. 4  illustrates a functional block diagram of a wireless power transmission system, in accordance with an exemplary embodiment. 
         FIG. 5  illustrates a block diagram of a wireless power transmitter, in accordance with exemplary embodiments. 
         FIGS. 6A-6B  illustrate a class-E amplifier including waveforms, in accordance with an exemplary embodiment. 
         FIG. 7  illustrates a circuit diagram of a loaded a asymmetric class-E amplifier, in accordance with an exemplary embodiment. 
         FIG. 8  illustrates a circuit diagram of a loaded symmetric class-E amplifier, in accordance with an exemplary embodiment. 
         FIG. 9  illustrates a circuit diagram of a loaded dual half bridge amplifier, in accordance with an exemplary embodiment. 
         FIG. 10  illustrates a circuit diagram of a filter and matching circuit including a waveform, in accordance with an exemplary embodiment. 
         FIGS. 11A and 11B  illustrate circuit diagrams of intermediate driver circuits, in accordance with exemplary embodiments. 
         FIG. 12  illustrates a circuit diagram of portions of a wireless power transmitter, in accordance with an exemplary embodiment. 
         FIG. 13  is a flowchart of a method for transmitting wireless power, in accordance with an exemplary embodiment. 
         FIG. 14  illustrates a circuit diagram of a wireless power receiver, in accordance with an exemplary embodiment. 
     
    
    
     DETAILED DESCRIPTION 
     The word “exemplary” is used herein to mean “serving as an example, instance, or illustration.” Any embodiment described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other embodiments. 
     The detailed description set forth below in connection with the appended drawings is intended as a description of exemplary embodiments of the present invention and is not intended to represent the only embodiments in which the present invention can be practiced. The term “exemplary” used throughout this description means “serving as an example, instance, or illustration,” and should not necessarily be construed as preferred or advantageous over other exemplary embodiments. The detailed description includes specific details for the purpose of providing a thorough understanding of the exemplary embodiments of the invention. It will be apparent to those skilled in the art that the exemplary embodiments of the invention may be practiced without these specific details. In some instances, well-known structures and devices are shown in block diagram form in order to avoid obscuring the novelty of the exemplary embodiments presented herein. 
     The term “wireless power” is used herein to mean any form of energy associated with electric fields, magnetic fields, electromagnetic fields, or otherwise that is transmitted from a transmitter to a receiver without the use of physical electromagnetic conductors. Power conversion in a system is described herein to wirelessly charge devices including, for example, mobile phones, cordless phones, iPod®, MP3 players, headsets, etc. Generally, one underlying principle of wireless energy transfer includes magnetic coupled resonance (i.e., resonant induction) using frequencies, for example, below 30 MHz. However, various frequencies may be employed including frequencies where license-exempt operation at relatively high radiation levels is permitted, for example, at either below 135 kHz (LF) or at 13.56 MHz (HF). At these frequencies normally used by Radio Frequency Identification (RFID) systems, systems must comply with interference and safety standards such as EN 300330 in Europe or FCC Part  15  norm in the United States. By way of illustration and not limitation, the abbreviations LF and HF are used herein where “LF” refers to f 0 =135 kHz and “HF” to refers to f 0 =13.56 MHz. 
       FIG. 1  illustrates wireless power transmission system  100 , in accordance with various exemplary embodiments. Input power  102  is provided to a transmitter  104  for generating a magnetic field  106  for providing energy transfer. A receiver  108  couples to the magnetic field  106  and generates an output power  110  for storing or consumption by a device (not shown) coupled to the output power  110 . Both the transmitter  104  and the receiver  108  are separated by a distance  112 . In one exemplary embodiment, transmitter  104  and receiver  108  are configured according to a mutual resonant relationship and when the resonant frequency of receiver  108  and the resonant frequency of transmitter  104  are matched, transmission losses between the transmitter  104  and the receiver  108  are minimal when the receiver  108  is located in the “near-field” of the magnetic field  106 . 
     Transmitter  104  further includes a transmit antenna  114  for providing a means for energy transmission and receiver  108  further includes a receive antenna  118  for providing a means for energy reception or coupling. The transmit and receive antennas are sized according to applications and devices to be associated therewith. As stated, an efficient energy transfer occurs by coupling a large portion of the energy in the near-field of the transmitting antenna to a receiving antenna rather than propagating most of the energy in an electromagnetic wave to the far-field. In this near-field, a coupling may be established between the transmit antenna  114  and the receive antenna  118 . The area around the antennas  114  and  118  where this near-field coupling may occur is referred to herein as a coupling-mode region. 
       FIG. 2  shows a simplified schematic diagram of a wireless power transmission system. The transmitter  104 , driven by input power  102 , includes an oscillator  122 , a power amplifier or power stage  124  and a filter and matching circuit  126 . The oscillator is configured to generate a desired frequency, which may be adjusted in response to adjustment signal  123 . The oscillator signal may be amplified by the power amplifier  124  with an amplification amount responsive to control signal  125 . The filter and matching circuit  126  may be included to filter harmonics or other unwanted frequencies and match the impedance of the transmitter  104  to the transmit antenna  114 . 
     An electronic device  120  includes the receiver  108  which may include a matching circuit  132  and a rectifier and switching circuit  134  to generate a DC power output to charge a battery  136  as shown in  FIG. 2  or power a device electronics (not shown) coupled to the receiver. The matching circuit  132  may be included to match the impedance of the receiver  108  to the receive antenna  118 . 
     As illustrated in  FIG. 3 , antennas used in exemplary embodiments may be configured as a “loop” antenna  150 , which may also be referred to herein as a “magnetic,” “resonant” or a “magnetic resonant” antenna. Loop antennas may be configured to include an air core or a physical core such as a ferrite core. Furthermore, an air core loop antenna allows the placement of other components within the core area. In addition, an air core loop may more readily enable placement of the receive antenna  118  ( FIG. 2 ) within a plane of the transmit antenna  114  ( FIG. 2 ) where the coupled-mode region of the transmit antenna  114  ( FIG. 2 ) may be more effective. 
     As stated, efficient transfer of energy between the transmitter  104  and receiver  108  occurs during matched or nearly matched resonance between the transmitter  104  and the receiver  108 . However, even when resonance between the transmitter  104  and receiver  108  are not matched, energy may be transferred at a lower efficiency. Transfer of energy occurs by coupling energy from the near-field of the transmitting antenna to the receiving antenna residing in the neighborhood where this near-field is established rather than propagating the energy from the transmitting antenna into free space. 
     The resonant frequency of the loop antennas is based on the inductance and capacitance. Inductance in a loop antenna is generally the inductance created by the loop, whereas, capacitance is generally added to the loop antenna&#39;s inductance to create a resonant structure at a desired resonant frequency. As a non-limiting example, capacitor  152  and capacitor  154  may be added to the antenna to create a resonant circuit that generates a sinusoidal or quasi-sinusoidal signal  156 . Accordingly, for larger diameter loop antennas, the size of capacitance needed to induce resonance decreases as the diameter or inductance of the loop increases. Furthermore, as the diameter of the loop antenna increases, the efficient energy transfer area of the near-field increases for “vicinity” coupled devices. Of course, other resonant circuits are possible. As another non-limiting example, a capacitor may be placed in parallel between the two terminals of the loop antenna. In addition, those of ordinary skill in the art will recognize that for transmit antennas the resonant signal  156  may be an input to the loop antenna  150 . 
     Exemplary embodiments of the invention include coupling power between two antennas that are in the near-fields of each other. As stated, the near-field is an area around the antenna in which electromagnetic fields exist but may not propagate or radiate away from the antenna. They are typically confined to a volume that is near the physical volume of the antenna. In the exemplary embodiments of the invention, antennas such as single and multi-turn loop antennas are used for both transmit (Tx) and receive (Rx) antenna systems since most of the environment possibly surrounding the antennas is dielectric and thus has less influence on a magnetic field compared to an electric field. Furthermore, antennas dominantly configured as “electric” antennas (e.g., dipoles and monopoles) or a combination of magnetic and electric antennas is also contemplated. 
     The Tx antenna can be operated at a frequency that is low enough and with an antenna size that is large enough to achieve good coupling efficiency (e.g., &gt;10%) to a small Rx antenna at significantly larger distances than allowed by far-field and inductive approaches mentioned earlier. If the Tx antenna is sized correctly, high coupling efficiencies (e.g., 30%) can be achieved when the Rx antenna on a host device is placed within a coupling-mode region (i.e., in the near-field or a strongly coupled regime) of the driven Tx loop antenna 
     As described herein, “proximity” coupling and “vicinity” coupling may require different matching approaches to adapt power source/sink to the antenna/coupling network. Moreover, the various exemplary embodiments provide system parameters, design targets, implementation variants, and specifications for both LF and HF applications and for the transmitter and receiver. Some of these parameters and specifications may vary, as required for example, to better match with a specific power conversion approach. System design parameters may include various priorities and tradeoffs. Specifically, transmitter and receiver subsystem considerations may include high transmission efficiency, low complexity of circuitry resulting in a low-cost implementation. 
       FIG. 4  illustrates a functional block diagram of a wireless power transmission system configured for direct field coupling between a transmitter and a receiver, in accordance with an exemplary embodiment. Wireless power transmission system  200  includes a transmitter  204  and a receiver  208 . Input power P TXin  is provided to transmitter  204  for generating a predominantly non-radiative field with direct field coupling  206  defined by coupling factor k for providing energy transfer. Receiver  208  directly couples to the non-radiative field  206  and generates an output power P RXout  for storing or consumption by a battery or load  236  coupled to the output port  210 . Both the transmitter  204  and the receiver  208  are separated by a distance. In one exemplary embodiment, transmitter  204  and receiver  208  are configured according to a mutual resonant relationship and when the resonant frequency, f 0 , of receiver  208  and the resonant frequency of transmitter  204  are matched, transmission losses between the transmitter  204  and the receiver  208  are minimal while the receiver  208  is located in the “near-field” of the radiated field generated by transmitter  204 . 
     Transmitter  204  further includes a transmit antenna  214  for providing a means for energy transmission and receiver  208  further includes a receive antenna  218  for providing a means for energy reception. Transmitter  204  further includes a transmit power conversion unit  220  at least partially function as an AC-to-AC converter. Receiver  208  further includes a receive power conversion unit  222  at least partially functioning as an AC-to-DC converter. 
     Various transmitter configurations are described herein which use capacitively loaded wire loops or multi-turn coils forming a resonant structure that is capable to efficiently couple energy from transmit antenna  214  to the receive antenna  218  via the magnetic field if both the transmit antenna  214  and receive antenna  218  are tuned to a common resonance frequency. Accordingly, highly efficient wireless charging of electronic devices (e.g. mobile phones) in a strongly coupled regime is described where transmit antenna  214  and receive antenna  218  are in close proximity resulting in coupling factors typically above 30%. Accordingly, various transmitter concepts comprised of a wire loop/coil antenna and a transmit power conversion unit are described herein. 
       FIG. 5  illustrates a block diagram of a wireless power transmitter, in accordance with exemplary embodiments. A transmitter  300  includes a transmit power conversion unit  302  and a transmit antenna  304 . Transmit power conversion unit  302  includes an amplifier  306 , an example of which is a class-E amplifier, that is used to drive transmit antenna  304 . A filter and matching circuit  308  provides load matching and/or filtering of the driving signal generated by the amplifier  306 . It is noted that the term “amplifier,” as used in relation to a class-E amplifier  306 , also corresponds to an “inverter,” “chopper,” or “power stage” because the amplification is highly non-linear and the main objective is to generate a substantially unmodulated signal for wireless power. 
     Transmit power conversion unit  302  further includes an oscillator  310  which generates a substantially unmodulated signal to intermediate driver  312  which in turn drives the amplifier  306 . Oscillator  310  may be implemented as a stable frequency source providing a square-wave signal with a 50% duty cycle. Intermediate driver  312  is configured to provide adequate drive for controlling transistors (e.g., MOSFETs) within the amplifier  306 . The different operating voltages required by the oscillator  310 , intermediate driver  312  and the amplifier  306  are generated by a DC/DC converter  314  in response to input voltage  316 . In one exemplary embodiment, the amplifier  306  receives oscillator signal  318  at a frequency of 13.56 MHz and amplifies the oscillator signal to a power level on the order of, for example, 7 watts. 
     The exemplary embodiment of  FIG. 5  provides an implementation based upon a reduced number of components and does not need additional circuitry to control the duty cycle due to the fixed duty cycle operation. Furthermore, the exemplary embodiment of  FIG. 5  can be implemented with a single transistor which results in a low harmonic content on the output signal due to a resonant load network needed for class-E operation. 
     Further by way of implementation, to design the class-E implementation of amplifier  306  and filter and matching circuitry  308 , the range of the antenna input impedance  322  and the load impedance  320  for the class-E operation of amplifier  306  need to be characterized. Further figures and description herein disclose measurements and modeling for determining those impedances. 
       FIG. 6A  illustrates an amplifier configured as a class-E amplifier, in accordance with an exemplary embodiment. An example of a transmitter configured with various amplifiers suitable for a wireless power transmitter operating at, for example, 13.56 MHz. A class-E amplifier  320  includes an active device switch  330 , a load network  332  and the load  334  illustrated as being a purely resistive load. The class-E amplifier  320  of  FIG. 6A  illustrates a single-ended class-E amplifier. 
     The load network  332 , including inductor L 1   340 , capacitor C 1   338 , capacitor C 2   336  and inductor L 2   334 , is used to shape the current and voltage waveform in a way that the active device switch  330  switches under zero-voltage and zero-current conditions. This heavily reduces switching losses since a major contributor to inefficiency is the power loss occurring in active device switch  330 . In addition, the parasitic capacitance (not shown) of the active device switch  330  (usually a FET) is used as a part of capacitor C 1   338  thus the negative influence of the parasitic capacitance is eliminated. 
       FIG. 6B  illustrates the resulting voltage and current waveforms of the active device switch  330  in a class-E configuration. At the switch-on instant (center of plot), the current and the voltage over the active device switch  330  is almost zero, leading to reduced switching losses. The same is true in the switch-off instant (end of plot) where the voltage rises only when the current is zero already. 
     Components for the class-E amplifier  306  may be determined according to the following formulae: 
                     L   1     =     10       ω   2     ·     C   1                 (   1   )                 L   2     =         Q   L     ·     R   Load       ω             (   2   )                 C   1     =     0.2     ω   ·     R   Load                 (   3   )                 C   2     =       1     ω   ·     Q   L     ·     R   Load         ·     (     1   +     1.11       Q   L     -   1.7879         )               (   4   )                 P   out     =     0.5768   ·         (       V   CC     -     V   CEsat       )     2       R   Load                 (   5   )                 V   CEpeak     =       3.563   ·     V   CC       -     2.562   ·     V   CEsat                 (   6   )               
(This formulae was given originally by the inventor of the class E amplifier, Nathan O. Sokal. Some reference should be given here (e.g. Sokal N. O., Sokal A. D., “Class E—a New Class of High Efficiency Tuned Single Ended Switching Power Amplifiers” IEEE Journal of Solid-State Circuits, Vol. SC-10, No. 3, June 1975)
 
     By way of implementation, the quality factor of the load network (Q L =ωL 2 /R Load ) has to be larger than 1.7879, otherwise capacitor C 2   336  becomes negative and a class-E configuration is inoperable. Furthermore, capacitor C 2   336  has to be larger than the collector-to-emitter capacitance (or drain-source capacitance) of the active device switch  330 . Accordingly, all the components of the load network are dependent of R Load . Since R Load  changes in the case of wireless power with the coupling factor (k) to the receiver, the load network probably has to be adjusted dynamically or needs to be designed for a good trade-off taking into account all operation conditions. 
     The class-E amplifier  320  of  FIG. 6A  can be adapted for wireless power transmission.  FIG. 7  illustrates a circuit diagram of an asymmetric class-E amplifier  350 , in accordance with an exemplary embodiment. At the transmit antenna input port, the coupling network comprised of the magnetically coupled transmit antenna and the loaded receive antenna can be represented in a first approximation by an L-R-circuit (equivalent resistance R_   eqv      362  and equivalent inductance L_   eqv      364  in  FIG. 7 ). Equivalent inductance L_   eqv      364  becomes a part of the load network (compare  FIG. 6A  component inductor L 2   334 ), and equivalent resistance R_   eqv      362  becomes the load resistance. Eventually, the inductor L_   eqv      364  is supplemented by an additional series inductor in order to increase the quality factor of the load network. The quality factor should be above 1.79, otherwise the class-E amplifier  350  cannot be designed properly as illustrated with respect of equations 1-6. 
     Supply voltage  352  provides the power from which the RF signals are generated based on the switching of control signal  356  which drives the active device switch  358 . A load network circuit includes inductor L 1   354 , capacitor C 1   360 , and capacitor C 2   368 . 
     The class-E amplifier  350  may generate harmonic content in the antenna current. To eliminate even order harmonics, a symmetric class-E stage may be used. Odd order harmonics need to be filtered with additional filtering circuitry.  FIG. 8  illustrates a circuit diagram of a class-E amplifier  400 , in accordance with an exemplary embodiment. A symmetric class-E amplifier  400  is an extension of the asymmetric class-E amplifier  350  ( FIG. 7 ) including a first class-E stage  416  and a second class-E stage  420  configured as a mirror of first class-E stage  416 . Signal generators  406 ,  426  operate at 180° phase shifted from each other and respectively drive switches  408 ,  428  at a 180° phase shifted waveforms resulting in a push-pull operation. 
     The two stages share the same load including equivalent resistance R_   eqv      412  and an equivalent inductance L_   eqv      414 . If equivalent resistance R_   eqv      412  and an equivalent inductance L_   eqv      414  remain unchanged compared to the asymmetric class-E amplifier  350  of  FIG. 7 , the capacitance of capacitors C 1 -C 4   410 ,  418 ,  430 ,  438  would have to be doubled in order to maintain Class E operation. This can be explained by the fact that effective inductance as seen per switch  408 ,  428  is half of equivalent inductance L_   eqv    (i.e., inductance L_   eqv    splits into two identical halves and is grounded at the symmetry point). 
     In the first class-E stage  416 , a supply voltage  402  provides the power from which the RF signals are generated based on the switching of control signal  406  which drives the active device switch  408 . A first load network circuit includes inductor L 1   404 , capacitor C 1   410 , and capacitor C 2   418 . In the second class-E stage  420 , a supply voltage  422  provides the power from which the RF signals are generated based on the switching of control signal  426  which drives the active device switch  428 . A second load network circuit includes inductor L 2   424 , capacitor C 3   430 , and capacitor C 4   438 . 
     The symmetric class-E amplifier  400  further eliminates even-order harmonic content in the current provided to the transmit antenna. Such even-order harmonic reduction reduces filtering circuitry that would otherwise be needed for supplementary second-harmonics filtering. Additionally, it can provide higher RF output power compared to the asymmetric class-E stage if both are operated from the same supply voltage. 
     The class-E amplifier desirably remains stable under different load conditions because various electronic devices or various receiver positions (with relation to the transmitter) of a receiver of an electronic device cause different load conditions. Changing the load condition on a class-E amplifier without adapting its load network will lead to a reduced efficiency and eventually higher stress on the active components. But depending on the type of load change, the impact could be smaller or larger. Various test cases have been simulated according to the component values listed in Table 1. 
                     TABLE 1                  Simulated test cases for class-E amplifiers       and their component values.                                             Case   1   2   3   4   5                                                         RL [Ω]   5   10   20   30   40           C1 [pF]   469   235   117   78   58           L1 [uH]   2.9   5.9   11.7   17.6   23.5           C2 [pF]   632   316   158   105   79           L2 [uH]   0.293   0.586   1.17   1.76   2.3           Vcc [V]   6.7   9.4   13.3   16.2   18.7                       (Note:           RL = target load resistance, Vcc = Supply voltage for the class-E amplifier)            
By now varying the load to become either capacitive or inductive, a desired operating region for the class-E power stage can be found. Circuit simulations have shown that the class-E power stage can be designed to operate efficiently on various loads such as produced by the different receiver coupling conditions that need to be supported.
 
     The component values and the required supply voltage were calculated using the formulae of Equations 1-6. The calculated values were optimized in the simulation to get the best possible efficiency with the target load (purely resistive). 
       FIG. 9  illustrates a circuit diagram of a dual half bridge amplifier, in accordance with an exemplary embodiment. A dual half bridge amplifier  450  drives a parallel tank circuit (not shown) and may be considered as the transformational dual circuit of a half bridge inverter driving a series tank circuit (not shown). The switching voltage and current waveforms are transformational duals to those of a class-D circuit. Compared to a class-E stage, the dual half bridge amplifier  450  does not need the additional shunt capacitors or any inductance to supplement the load network. As opposed to the classical half bridge topology, the dual half bridge provides low dV/dt voltage waveforms and switching is ideally performed at zero voltage instants. Switch transistors junction capacitance (e.g. drain-source capacitance of a FET) may be considered integral part of capacitance needed to achieve resonance in the antenna parallel tank circuit. So there is no abrupt charge and discharge of junction capacitances when switches open and close at zero voltage instants. However, the dual half bridge amplifier may be more susceptible to variations of equivalent inductance L_   eqv    thus resonant frequency of the parallel tank circuit as the result of any changes in the wireless power link (coupling network) as shown below. In order to achieve or maintain zero voltage switching, switch voltage needs to be phase aligned to switch current 
     In a first order approximation, the coupling network consisting of the magnetically coupled transmit and the loaded receive antenna may be represented at its input port by an L-R series circuit (L_   eqv      470 , R_   eqv      468 ). A parallel capacitor C 1    466  is added to compensate for the inductive part of the antenna. Proper design and adjustment of capacitor C 1    466  is important as it results in a highly efficient operation of the dual half bridge because any non-compensated reactive part in the load causes a phase-shift between the switch voltage and the switch current making it infeasible to switch transistors in a lossless mode. Since equivalent inductor L_   eqv      470  and equivalent resistance R_   eqv      468  vary with the coupling to the receiver, capacitor C 1    466  should be adjusted dynamically, if high efficiency is to be maintained in all coupling and loading conditions. 
     A supply voltage  460  provides the power from which the RF signals are generated based on the switching of control signals  452  and  454  which respectively drive the active device switches  4456  and  458 . Chokes L 2   462  and L 3   464  are used to provide a substantially constant current to the active device switches or the load and to filter RF currents from the supply voltage  460  (compare L 1   340  in  FIG. 6 a   ). The dual half bridge amplifier  450  may also be configurable in a receiver configured to operate as a synchronous rectifier operating in the positive VI quadrant, as part of a wireless power receiver. 
       FIG. 10  illustrates a circuit diagram of a filter and matching circuit  308  of  FIG. 5  and the respective frequency response. The filter and matching circuit  308 , also known as a “resonant transformer” or “L-section”, provides an effective approach to realize narrowband matching and a certain additional filtering effect. The impedance gradient results in the filter and matching circuit  308  being well suited to be combined with amplifier  306  ( FIG. 5 ) configured as a class-E amplifier, since for the harmonics (e.g. at 27.12 MHz and 40.68 MHz), the filter and matching circuit  308  represents a high impedance. The bandwidth, or Q-factor, of the filter and matching circuit  308  is related to the ratio of resistance R 1   446  to resistance R 2   448 . A higher impedance-ratio leads to a narrower bandwidth and therefore to a higher filtering effect. A matching network with a high impedance ratio results when a class-E amplifier that is designed for a low target load impedance (e.g. 8Ω) is matched to an antenna  304  that is a parallel tank ( FIG. 5 ) typically presenting a high input impedance. In case of a charging pad antenna this impedance may be 700Ω when loaded with a receiver. This approach seems particular interesting for class E amplifiers that need to operate from low DC supply voltages and that typically perform near optimum in terms of both RF power output and efficiency when designed for low target load impedances. 
       FIG. 11A  and  FIG. 11B  illustrate circuit diagrams of intermediate driver circuits, in accordance with exemplary embodiments. As illustrated in  FIG. 5 , intermediate driver  312  drives the amplifier  306 . Selection of intermediate driver  312  contributes to the efficiency of the transmitter  300  since power consumption by the intermediate driver  312  reduces the overall efficiency while the output signal of the intermediate driver  312  influences the switching behavior of the amplifier  306  thereby affecting the efficiency of an amplifier  306  configured as a class-E amplifier. 
       FIG. 11A  and  FIG. 11B  illustrated two different intermediate driver types.  FIG. 11A  illustrates a resonant-type intermediate driver  312 ′ which utilizes energy stored in the gate capacitance of the transistor  482  (e.g., MOSFET) by adding an inductor  480  to build a series tank circuit. Such an approach appears to perform adequately for higher power levels and lower frequencies, but for power levels below, for example, 10 Watts at 13.56 MHz, the additional circuitry (inductor, diodes and more complex control signals) for a resonant gate driver may incur additional complexity. 
       FIG. 11B  illustrates a nonresonant-type intermediate driver  312 ″ which exhibits a comparable efficiency to the resonant-type intermediate driver  312 ′ of  FIG. 11A . By way of example, a nonresonant-type intermediate driver  312 ′ may be configured as push-pull gate drivers including a totem-pole output stage with an N-channel transistor (e.g., MOSFET) and a P-channel transistor (e.g., MOSFET) as illustrated in  FIG. 11B . By way of implementation, to achieve a high efficiency with a push-pull intermediate driver, the intermediate driver should provide low r DSon  values, fast switching speeds and a low inductance design to prevent ringing. To reduce resistive losses in the driver, several push-pull stages can be used in parallel. 
     A description has been provided for a wireless power transmitter configured to include an amplifier configured as a class-E amplifier. Various implementation considerations include the realization that generally, for a given volume, low inductance values can be realized with a higher quality factor than high inductance values. Furthermore with reference to  FIG. 5 , the oscillator  310 , intermediate driver  312  and other auxiliary components (e.g., controllers) in wireless power transmitter  300  desirably operate from the same auxiliary voltage since each additional DC-DC conversion introduces a power loss and requires additional volume within the electronic device. Further design considerations include operation of the active device switch (e.g., MOSFET) at high drain voltages (e.g. 75 V for a 100 V type) and low drain currents since the R DSon  of the used MOSFET types may be quite high, leading to higher losses with increased drain currents (could change with future semiconductors). Consequently, the target load impedance of the class-E amplifier (set by the L-section matching circuit) is desirably in a range where a good trade-off between high efficiency and high power output for given supply voltage is achieved. Optimized values, in accordance with exemplary embodiments, are in the range of 5Ω to 15Ω. 
       FIG. 12  illustrates a circuit diagram of portions of a wireless power transmitter, in accordance with an exemplary embodiment. Class-E amplifier  306 ′ of  FIG. 12  illustrates a partial circuit diagram of the class-E amplifier of  FIG. 5  and as further detailed in  FIG. 6A ,  FIG. 7  and  FIG. 8 . Specifically, class-E amplifier  306 ′ illustrates the active device switch  330 , the capacitor C 2   336 , and the inductor L 2   334 .  FIG. 12  also illustrates filter and matching circuit  308  of  FIG. 5  and as further detailed with respect to  FIG. 10 . The filter and matching circuit  308  of  FIG. 5  includes the inductor LF  472  and the capacitor CF  474 . 
       FIG. 12  illustrates that the series-configured arrangement of the inductor L 2   334  of the class-E amplifier  306 ′ and the inductor LF  472  of the filter and matching circuit  308 . Accordingly, amplifier inductor L 2   334  and filtering inductor LF  472  may be combined into a single element inductor  476 . Wireless power transmitter  300  further includes an antenna  304  configured as a tank circuit including an antenna capacitor CA  478  and an antenna inductor LA  480  that will generally include losses which may be modeled by a series loss resistance (not shown in  FIG. 12 ). Additionally, circuit elements  484  and  486  are included to model self capacitance (e.g. in case of a multi-turn loop antenna) and losses e.g. due to the electric stray field and presence of lossy dielectric materials. The magnetic coupling between antenna inductor LA  480  and the receive antenna is not shown in  FIG. 12 .  FIG. 12  illustrates the parallel-configured arrangement of the filtering capacitor CF  474  and the antenna capacitor CA  478 . Accordingly, the filtering capacitor CF  474  and antenna capacitor CA  478  may be combined into a single element capacitor  482 . 
     Therefore,  FIG. 12  illustrates how a transmit antenna may be efficiently driven by a class-E amplifier with the components being reduced to a single inductor  476 , a single capacitor  482  and a loop antenna inductor  480 . Accordingly, a wireless power transmitter has been described which results in a low bill of materials due to the use of an amplifier, matching filter, a transmit antenna combination that allows for a combination of reactive components. Furthermore, the selection of an amplifier, a matching filter and a transmit antenna allows for a combination of reactive components also resulting in a reduced number of components. Also, in an exemplary embodiment utilizing a symmetric class-E amplifier, the second harmonics of the antenna current are canceled. 
       FIG. 13  illustrates a flowchart of a method for transmitting wireless power, in accordance with an exemplary embodiment. Method  500  for transmitting wireless power is supported by the various structures and circuits described herein. Method  500  includes a step  502  for driving a transmit antenna from an amplifier through a matching circuit. Method  500  further includes a step  504  for resonating the transmit antenna according to a transmit antenna capacitance realized in a capacitor shared with the matching circuit. 
     By way of an example of a wireless power receiver,  FIG. 14  illustrates a circuit diagram of a wireless power receiver, in accordance with an exemplary embodiment. Wireless power receiver  608  including a resonant receive antenna  618 , including inductive loop L 2    632  and capacitor C 2    634 , and a passive double diode full wave rectifier circuit  600 , in accordance with an exemplary embodiment. Rectifier circuit  600  includes diode D 21    628  and diode D 22    630 . Rectifier circuit  600  further includes a high frequency (HF) choke L HFC    624  and a high frequency (HF) block capacitor C HFB    626 . The DC path is closed via the antenna loop. HF choke  624  acts as current sink and with a diode conduction cycle D of 50%, the voltage transformation factor M is 0.5. Furthermore, the input impedance as seen at terminals A 2 , A 2 ′ at a fundamental frequency is approximately 4 times the load resistance R L    636 . 
     Proper selection of diodes for rectifier circuit  600  may reduce circuit losses and increase overall efficiency. For rectification efficiency, diodes may be selected based upon various parameters including peak repetitive reverse voltage (V RRM ), average rectified forward current (I 0 ), maximum instantaneous forward voltage (V F ), and junction capacitance (C j ). V RRM  and I 0  are maximum ratings of the diode, whereas V F  and C j  are characteristic values which influence the efficiency of the rectifier. 
     Various diodes were tested and voltage, current and instantaneous power of each diode was calculated during the simulation to characterize the switching behavior of each type. Different switching behavior of the tested diodes were observed with the diode with the largest Cj (MBRA340T3) showing the worst switching behavior but exhibiting the smallest ON-state loss due to the reduced forward voltage. For the various diodes types tested, the ON-state loss was dominant. Specifically, the switching loss varies with the junction capacitance and the ON-state loss varies with the forward voltage. Accordingly, the total loss is dependent on the ratio of C j  and U F  and the operating point of the diode, which is dependent on load resistance R L    636 . 
     A configuration of two parallel PMEG4010EH diodes proved to be the best option since the switching loss of this diode type is very small and the ON-state loss is reduced due to the parallel configuration. The rectifier diodes may be implemented as double diodes (second diodes shown in phantom) to reduce the conduction losses. The current splits up equally on both diodes thus changing the operation point of each diode compared to a single diode solution. It was also observed that a single MBRS2040LT3 performed similarly well because the forward voltage was significantly lower compared to the PMEG4010EH and the switching loss was still reasonable. Accordingly for one exemplary embodiment, a diode with a junction capacitance of about 50 pF and a forward voltage of about 380 mV@ 1 A is an acceptable choice. 
     Those of skill in the art would understand that control information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof. 
     Those of skill would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, and controlled by computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented and controlled as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the exemplary embodiments of the invention. 
     The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be controlled with a general purpose processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration. 
     The control steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in Random Access Memory (RAM), flash memory, Read Only Memory (ROM), Electrically Programmable ROM (EPROM), Electrically Erasable Programmable ROM (EEPROM), registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such that the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a user terminal. In the alternative, the processor and the storage medium may reside as discrete components in a user terminal. 
     In one or more exemplary embodiments, the control functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code in the form of instructions or data structures and that can be accessed by a computer. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media. 
     The previous description of the disclosed exemplary embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these exemplary embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.