Patent Publication Number: US-10781785-B2

Title: Circuit and method for soft shutdown of a coil

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a continuation of U.S. patent application Ser. No. 16/045,469 filed on Jul. 25, 2018, which is incorporated by reference herein in its entirety. 
    
    
     FIELD OF THE DISCLOSURE 
     The present disclosure relates generally to an ignition system and more specifically, to a circuit and method for shutting down (i.e., discharging) an ignition coil gradually (i.e., softly) to prevent an unwanted spark. 
     BACKGROUND 
     Engines (e.g., vehicle engines) may have an ignition system for starting. Generally speaking, the ignition system includes a battery that is connected to a primary coil of a step-up transformer and an ignitor switch. When the ignitor switch is closed, current flows from the battery to the primary coil. After some period, the switch is opened and the current from the battery to the primary coil is abruptly stopped. The abrupt change in current through the primary coil causes a large transient voltage across the primary coil. The transient voltage is stepped up through the transformer and because the secondary coil is in series with a spark gap, the voltage at the secondary coil produces a spark at the spark gap. In some situations, however, it is desirable to turn off the current in the primary coil (i.e., shutdown, de-energize) without creating a spark at the spark gap. 
     SUMMARY 
     In one general aspect, the present disclosure generally describes a circuit. The circuit includes a controller that is configured in a control loop with a current sensor and a transistor (e.g., an insulated-gate bipolar transistor) that is configured to adjust a voltage at a terminal of the transistor to reduce a current through the transistor. The controller has an open loop gain determine be a resistance of a variable feedback resistor. The circuit further includes a signal generated that is coupled to the controller. The signal generator generates a ramp signal to control a reference level of the controller so that the current is reduced gradually over a period. The controller also generates the ramp signal to control the resistance of the variable feedback resistor over the period to reduce an open loop gain of the controller. 
     In one possible implementation the period includes a high current region and a low current region. Further, the current can be ramped linearly downward according to the reference voltage when the current is in the high current region but may deviate from the reference voltage when the current is in the low current region. This deviation can increase a phase margin in the low current region, which may correspond to an increase in stability. 
     In another general aspect, the present disclosure generally describes a method. The method includes sensing a current through a transistor that is configured in a control loop with a current sensor and a controller. The method further includes generating, using a signal generator, a ramp signal that decreases linearly over a period. The method further includes reducing a reference level of the controller according to the ramp signal. The method further includes adjusting, using the controller, a voltage at a terminal of the transistor to reduce the current through the transistor over the period. The adjustment is based on a comparison between the reduced reference level and the sensed current level. The method further includes controlling a resistance of a variable feedback resistor of the controller to reduce an open loop gain of the controller during the period. 
     In one possible implementation of the method, adjusting (using the controller) the voltage at the terminal of the transistor to reduce the current through the transistor over the period can include reducing the current through the transistor according to a first profile that matches a profile of the ramp signal in a high current region of the period. Further, this operation can also include reducing the current through the transistor according to a second profile that does not match (i.e., deviates from) the profile of the ramp signal in a low current region of the period. The deviation of the second profile can increase a phase margin of the controller in the low current region, which can correspond to an increase in stability of the controller (e.g., in the low current region). 
     The foregoing illustrative summary, as well as other exemplary objectives and/or advantages of the disclosure, and the manner in which the same are accomplished, are further explained within the following detailed description and its accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a simplified schematic of a spark ignitor circuit including an ignition coil according to an embodiment of the present disclosure. 
         FIG. 2A  is a graph of a coil current versus time illustrating a hard shutdown profile. 
         FIG. 2B  is a graph of coil current versus time illustrating a soft shutdown profile. 
         FIG. 3  is a block diagram of a current limiter for controlling a current in a coil according to an embodiment of the present disclosure. 
         FIG. 4  is a schematic of a current limiter circuit with a variable feedback resistance for the soft shutdown of a coil according to an embodiment of the present disclosure. 
         FIG. 5  are graphs of feedback resistance (Rfb), open loop gain, reference voltage (Vref), sensed voltage (Vsns), and coil current of the current limiter circuit of  FIG. 4  during a soft shutdown period. 
         FIG. 6  is a detailed schematic of a current limiter circuit with variable feedback resistance according to a possible embodiment of the present disclosure. 
         FIG. 7  is a section of the schematic of  FIG. 6  showing the variable feed back resistor according to a possible embodiment of the present disclosure. 
         FIG. 8  is a detailed schematic of a current limiter circuit with variable feedback resistance according to another possible embodiment of the present disclosure. 
         FIG. 9  is a schematic showing a possible variation of sections of the current limiter circuit with variable feedback resistance of  FIG. 8 . 
         FIG. 10A  is a graph of feedback resistance provided by the variable feedback resistor of  FIG. 7  during a shutdown period as the soft shutdown current is increased. 
         FIG. 10B  is a graph of gate voltages at each transistor controlling a resistor in the variable feedback resistor of  FIG. 7  during a shutdown period as the soft shutdown control current is increased. 
         FIG. 11  is a graph of coil current versus control current during soft shutdown for a current limiter circuit having a fixed feedback resistance and for a current limiter circuit having a variable feedback resistance. 
         FIG. 12  is a plot of gain and phase versus frequency illustrating an equivalent phase margins in a high current region (12 A) of a soft shutdown for both fixed and variable feedback resistance embodiments. 
         FIG. 13  is a plot of gain and phase versus frequency illustrating different phase margins in a low current region (1 A) of a soft shutdown for fixed and variable feedback resistance embodiments. 
         FIG. 14  is a schematic of a circuit in which a loop gain is controllable over operating conditions to adjust the accuracy and stability of the output. 
     
    
    
     DETAILED DESCRIPTION 
     The present disclosure describes a circuit and method for soft shutdown of an ignition circuit (i.e., igniter circuit) that utilize a negative feedback loop with a controlled open loop gain to increase a phase margin as a coil current is reduced, which insures stability as coil currents are reduced. While variations may exists, the ignition circuit can operate in a vehicle environment (i.e., to start an engine). An example ignition circuit that can be used in the implementations described herein is shown in  FIG. 1 . 
       FIG. 1  illustrates an ignition circuit  100  includes a battery  130  that supplies a coil current to a primary winding (i.e., primary coil) of a transformer  110  when a switch  140  is closed (e.g., turned ON). When switch  140  is opened (e.g., turned OFF) the coil current is stopped abruptly and a large voltage (e.g., 400 volts (V)) is created across the primary coil due to the inductance of the primary coil (i.e., V=L di/dt). This voltage at the primary coil is transformed by the transformer  110  to a higher voltage at the secondary coil (e.g., 30 kilovolts (KV)) due to a large windings ratio, N 2 /N 1  (e.g., 50-100). The large voltage appears at a spark gap (e.g., spark plug gap)  120 , and is large enough to overcome the resistance of the gap to create a spark. 
     In some cases, the coil current is brought to a high level for a period of time that is larger than expected or desired. For example, if a high coil current remains for a long period (e.g., greater than a timeout period, Ton) without a spark, then damage (e.g., overheating) to an ignition system could result. Accordingly, the coil current should be shut off before damage occurs. In practice, a timeout period may be defined and if a coil current remains high for a period greater than a timeout period, Ton, then the coil current is shut off. 
     As shown in  FIGS. 2A and 2B , there are two ways to shut off the coil current after the timeout period, Ton. The first way, shown in  FIG. 2A , is a hard shutdown in which the coil current is abruptly shut off (e.g., by opening a switch) after the time out period Ton. As discussed, this method produces a spark. The expiration of the time out period, Ton, however, may not be correlated with the operation/state of the engine. As a result, a spark from a hard shutdown may occur regardless of a cylinder position and an undesirable combustion may occur as a result of the abnormal spark. 
     In order to de-energize the coil and shut off the ignition system without a spark the coil current can be reduced gradually.  FIG. 2B  is a graph of a coil current associated with a simple soft shutdown. After the expiration of the timeout period, Ton, the coil current is brought to zero gradually over some shutdown (i.e., shutoff) period, Toff. Because the coil current is shutdown gradually, the voltage created across the primary coil is never made large enough to overcome the resistance at the spark gap. Thus, no spark is created. The times for Ton and Toff are selectable based on the application and the devices used. Additionally, the time for the primary coil to ramp up (i.e., charge) to steady state is not necessarily related to Toff. The present disclosure describes circuits and methods that can generally accommodate either the hard shutdown of  FIG. 2A  or the soft shutdown of  FIG. 2B , but as will be described, can also control the coil current during shut down in a way that balances control accuracy with control stability as the coil current levels are reduced. 
     A block diagram of the disclosed current limiter circuit  300  for controlling the coil current during shutdown is shown in  FIG. 3 . In the circuit, the coil current level is sensed by a shunt resistor  310  and measured by a current sense circuit  320 . The measured current (i.e., sensed current level) is then compared to a reference current level (i.e., current limit level)  315  at a current limit controller  330  (i.e., controller). Based on the comparison, a difference signal (i.e., error signal) is obtained, amplified, and inverted to create an IGBT gate control signal. The IGBT gate control signal, with the aid of the IGBT gate driver  340 , control the operating point of the IGBT  350  to raise/lower the coil current to according to the error signal. The gate driver  340  amplifies the current supplied to the gate of the transistor based on the gate voltage to charge/discharge the gate capacitance quickly. The gate voltage on the IGBT controls the coil current so that it never exceeds the reference current limit level (Icl)  315 . This control may be utilized to perform a soft shutdown of the coil current. For example, if the reference current limit level (Icl)  315  of the control circuit  300  (i.e., current limiter circuit) is reduced gradually over a time, Toff, then the coil current will be reduced gradually as well and a soft shutdown discharge of the coil will be achieved. 
     A schematic of a current limiter circuit  400  according to an embodiment of the present disclosure is shown in  FIG. 4 . The current limiter circuit  400  is coupled to an ignition coil  401 . In operation, a feedback voltage (Vsns)  415 , created by a coil current (Icl) across the shunt resistor (Rsns)  310 , is compared by a differential amplifier  420  to a reference voltage, Vref  425 . The reference voltage, Vref,  425  results from a variable voltage source  405  controlled by a ramp signal  403  created by a signal generator  410 . The resulting difference signal  435  is amplified by an inverting amplifier  440  with a gain controlled by a variable feedback resistor (Rfb)  450 . The output of the inverting amplifier  440  is fed to an IGBT gate driver  340  to control the gate of the IGBT  350 , which adjusts the current limit of the primary coil in the ignition coil circuit  401  (i.e., the coil current) to reduce (e.g., minimize) the difference signal  435 . 
     In other words, the circuit includes a controller  430  that is configured in a control loop with the current sensor  310  and a transistor  350 . The controller  430  is configured to adjust the voltage at the gate of the transistor  350  to reduce the difference between the sensed voltage level (Vsns) and a reference voltage level (Vref). In this configuration, the controller  430  has an open loop gain determined by parameters that include the resistance of the variable feedback resistor  450 . 
     The feedback process described can be repeated (in real time) until the coil current (i.e., Vref) is brought to zero.  FIG. 5  includes graphs of the sensed voltage, Vsns  415 , (dotted line) and the reference voltage, Vref  425 , (solid line) during a soft shutdown period  515 . As can be seen, the coil current (Icl) generally follows the reference voltage as it is ramped down during the soft shutdown period  515 . 
     As mention previously, the current limiter circuit  400  relies on a negative feedback control system. The stability of the control system may change as the controlled coil current changes. An aspect of the present disclosure is the recognition that to prevent instability as the coil current (Icl) is reduced, the open loop gain of the current limiter circuit  400  can also be reduced to increase phase margin (i.e., stability) of the current limiter circuit  400 . Another aspect of the present disclosure is the recognition that the added stability comes at the expense of control accuracy and that the trade of accuracy for stability is best suited for a low current region of a soft shutdown profile where control accuracy is less important. 
     Reducing the feedback resistance (Rfb)  450  can reduce the open loop gain, which can improve the phase margin (i.e., stability) of the current limiter circuit  400 . As mentioned, the open loop gain reduction can lead to a loss of the control accuracy due to a variety of error sources (e.g., temperature, process, IGBT gate-emitter voltage, etc.). In other words, reducing the feedback resistance (Rfb)  450  may cause the coil current to deviate from the soft shutdown profile as prescribed by Vref. This deviation could lead to unwanted sparking in a high current region  505 . Thus, the disclosed current limiter circuit controls the feedback resistance (i.e., open loop gain) to strike a balance between stability in a low current region  500  accuracy in a high current region  500 . The exact balance may vary based on the application and operating points. 
     The circuits and methods described herein offer the advantage of an improved phase margin (i.e., stability) in low a current region  500  (i.e., where stability can be problematic), while maintaining current limit accuracy in a high current region  505  (i.e., where following a soft shutdown profile is important). 
     In the current limiter circuit  400 , the controller  430  creates a difference signal  435  (i.e., error signal) that is amplified by an inverting amplifier  440  with a gain controlled by a variable feedback resistor (Rfb)  450 . The value (i.e., resistance) of the variable feedback resistor  450  is controlled by the signal generator  410  according to a ramp signal. The output of the inverting amplifier  440  is fed to an IGBT gate driver that drives the gate of the IGBT to adjust the current limit of the primary coil in the ignition coil circuit  401  to reduce the difference signal. 
     Various profiles (i.e., soft shutdown profiles) of measured parameters in the current limiter circuit  400  are graphed in  FIG. 5 . As shown, Vref,  425  can be ramped linearly down from a voltage to zero voltage over a soft shut down period  515 . The precise amplitude and timing profile of Vref  425  during the soft shutdown period  515  are determined by the ramp signal  403  from a signal generator  410  and may be selected based on the application (e.g., ignition coil parameters). In addition to controlling Vref, the ramp signal  403  from the signal generator  410  also controls the resistance (Rfb) of the variable feedback resistor  450 . As shown in  FIG. 5 , the resistance, Rfb, is gradually reduced based during the shutdown period  515 . The reduction of Rfb may or may not decrease linearly, as Vref does. The exact profile of Rfb may be selected to achieve a desired balance between phase margin (i.e., stability) in the low current region  500  of the soft shutdown period and accuracy in a region (or regions) where it is more important, such as a high current region  505  of a soft shutdown period and/or a current limit period before the soft shutdown period. Additionally, the maximum and minimum feedback resistance Rfb values may be adjusted with this balance in mind as well. 
     The open loop gain of the circuit  400  is related to the feedback resistance (Rfb) of the inverting amplifier  440  and thus, decreases based on the decreasing value of Rfb. As mentioned previously, the reduction in the open loop gain in the low current region  500  maintains phase margin (i.e., stability) at the cost of control accuracy of the coil current. As the open loop gain is reduced, the coil current (and therefore Vsns) follow a soft shutdown profile that corresponds to Vref less accurately in the low current region  500  of the soft shutdown period than in the high current region  505  of the soft shutdown period. 
       FIG. 6  schematically depicts the circuit details of a possible embodiment of the variable feedback resistance current limiter circuit coupled to an ignition coil. In the circuit, a signal generator  610  includes a voltage ramp wave fed to a voltage to current converter  615  to create a corresponding soft shutdown control current, Issd. The current, Issd, is coupled out of the signal generator using a current mirror. 
     At a first input of the differential amplifier  620 , the current (Io-Issd) flows to ground through a resistor, Rref, to produce a voltage, Vref at the differential amplifier&#39;s first input. The reference voltage, Vref, follows a soft shutdown profile. In other words, the voltage, Vref, ramps down from a high voltage (i.e., Io-Issd is large) to a low voltage (i.e., Io-Issd is small). At a second input of the differential amplifier,  620 , the coil current (Icl) flows to ground through a sensing resistor (Rsns)  310  thereby creating a sensed voltage (Vsns) corresponding to the coil current. The output of the differential amplifier  620  corresponds to the difference voltage (Vsns-Vref). 
     The difference voltage (Vsns-Vref) is coupled to an input of an inverting amplifier  640 . The inverting amplifier may be embodied variously. The inverting amplifier shown in  FIG. 6  is a transistor (e.g., n-type MOSFET) with a gain that is determined by a feedback resistance (Rfb) between the drain and the gate of transistor. A high feedback resistance corresponds to a high gain and a low feedback resistance corresponds to a low gain (e.g., zero resistance corresponds to zero gain-diode). 
     After the inverting amplifier  640 , the inverted and amplified output voltage is applied at the gate of the IGBT  350  and effectively pulls down or pulls up the output of the IGBT driver to control the coil current according to the amplified and inverted voltage difference. For example, if the difference Vsns-Vref is positive then the coil current is reduced. Thus, if Vref is a ramp voltage that decreases to zero of a period (Toff) then the coil current will, ideally, follow the same ramp profile to zero over the same shutdown period (Toff). It should be noted that the present disclosure envisions that signals other than ramp waves could be used to control the shutdown of the coil. 
     An aspect of the disclosure is the open loop gain of the control loop (i.e., the gain of the inverting amplifier) is lower as the coil current reaches a low current region of a soft shutdown profile. The open loop gain is reduced by decreasing the feedback resistance value Rfb of the variable feedback resistor  450 . The variable feedback resistor  450  may be embodied and controlled in various ways and while the present disclosure presents several possible embodiments, variations (e.g., number of resistors, values of each resistor, etc.) are understood to be within the cope the of the present disclosure.  FIG. 7  is a detailed schematic of the variable feedback resistor  630  shown in the current limiter circuit of  FIG. 6 . 
     As shown in  FIG. 7 , the variable feedback resistor  630  of  FIG. 6  consists of a bank of series connected resistors (R 1 -R 8 )  710  that are each connected in parallel with one transistor in of a bank of transistors (M 11 -M 18 )  720 . When a particular transistor in the bank of transistor is turned ON, the resistor that is in parallel with the particular transistor is shorted to reduce the overall series resistance of the variable feedback resistor. For example, R 1  is connected between the source and drain of M 11 . If the voltage at the gate of M 11  turns the transistor ON then the impedance between the source and drain of M 11  is much smaller than the resistance of R 1 . Accordingly R 1  is shorted and the overall feedback resistance becomes R 2 +R 3 +R 4 +R 5 +R 6 +R 7 +R 8 +R 9 . The additional resistor, R 9 , may be included without any corresponding transistor to set the minimum feedback resistance (i.e., open loop gain) of the amplifier. 
     To gradually reduce the feedback resistance Rfb, the transistors can be successively turned ON over the soft shutdown period. There are a variety of ways to control the transistors  720  according to the soft shutdown profile. While the disclosure presents several possible embodiments of transistor control circuits, these do not comprise an exhaustive list and other possible embodiments or variations (e.g., transistor type, connection configuration, etc.) are considered within the scope of the present disclosure. 
     One possible embodiment of a transistor control circuit  730  is shown in  FIG. 7 . As shown, the circuit receives the ramped soft shut down control current, Issd, from the signal generator (e.g., via a current mirror in the signal generator). The transistor control circuit  730  includes a voltage divider (R 11 -R 18 ) that is tapped at different voltages by a terminal (e.g., the gate, the base, etc.) of each of the transistors (M 11 -M 18 ) in the bank of transistors  720 . The tap points provide voltages that increase by the voltage drop across each resistor in the voltage divider. The gate-drain connected transistors (M 1 , M 2 , and M 3 ) operate as series connected diodes are used as part of the circuit because a current, Issd, is received at the voltage divider instead of a voltage. 
       FIG. 8  schematically depicts the circuit details of another possible embodiment of the variable feedback resistance current limiter circuit coupled to an ignition coil. In the circuit, a voltage ramp wave from the signal generator  830  is coupled to a transistor control circuit via a buffer amp  810 . The transistor control circuit  820  includes to a bank of series connected resistors acting as a voltage divider to produce the voltage taps that control the transistors to sequentially turn on as the ramp wave signal progresses from low to high. Like the previous embodiment, the voltage divider is tapped at different voltages by the gates of each of the transistors in the bank of transistors. The tap points provide voltages that increase by the voltage drop across each resistor in the voltage divider. Accordingly, the transistors may be turned on in succession as the ramp wave voltage is increased. 
     A possible variation to the circuit of  FIG. 8  is shown in  FIG. 9 . Here, the voltage ramp wave is obtained from a node in the signal generator circuit, but additional variations are possible. For example, thus far it has been assumed that the signal generator provides a ramp signal for control of both the reference voltage (Vref) and the feedback resistance (Rfb) together. The control of the feedback resistance, however, may be controlled separately. For example, a dedicated controller may be used to provide the signal necessary to control the transistors in the variable feedback resistor during soft shutdown independently of the control of the coil current. In general, various adaptations and modifications of the variable feedback resistance current limiter circuits described above can be configured without departing from the scope and spirit disclosure. In other words, as long as Rfb is reduced during Toff, phase margin for low coil currents can be maintained. The particular shape and timing of the resistance reduction may be practiced variously within the scope of the present disclosure. 
     The particular reduction of the resistance of the variable feedback resistor of  FIG. 7  during a shutdown transition (i.e., as the soft shutdown current, Issd, is increased) is shown in  FIG. 10A . As shown, the feedback resistance starts as the total of all resistance in the bank of resistors  710  but steadily falls as the gate voltages of each transistor is raised to turn on each of the transistors in succession. As each transistor is turned on, resistors in the bank of resistors are shorted until only R 9  remains. 
       FIG. 10B  is a plot of the gate voltages versus time during a shutdown transition (i.e., as the soft shutdown current, Issd, is increased) for the bank of transistors  720  of  FIG. 7 . In this case (as in  FIG. 10A ), Issd is linear with time (i.e., is based on a linear ramp wave) thus the x-axis of the plot may also be considered as time. As can be observed, the transistors each reached a threshold voltage to turn ON at a different time. The turn ON time is determined by the voltage divider circuit and the ramp signal applied thereto. 
     The resistance transition shown in  FIG. 10A  can be controlled by the divider ratio of the voltage divider circuit and the number of transistors. In other words, the variable resistor may have any number of resistors and transistors, and the resistors in the bank of resistors may the same or different values in order to control the feedback resistance profile in  FIG. 10A . The minimum resistance can be set based on the resistance (e.g., R 9 ) with no corresponding transistor and the transistors (M 11 - 18 ) act as analog switches and respond more smoothly to the transition of Rfb than logical switches. 
       FIG. 11  illustrates a comparison between (i) the coil current of the current limiter circuit when the feedback resistance is varied (e.g., by a ramp signal applied to the variable feedback resistor) during a soft shutdown period and (ii) the coil current of the current limiter circuit when feedback resistance is not varied (e.g., by no ramp signal or a constant signal applied to the variable feedback resistor). As shown, the coil current (i.e., dotted line) corresponds linearly to a control current (Issd) from the signal generator for the fixed feedback resistance case for both high coil currents (e.g., approximately 12 amps) and low currents (e.g., approximately 1 amp). The coil current (i.e., solid line) for the variable feedback resistance case, however, corresponds linearly to the control current for high currents (e.g., approximately 12 amps) but does not correspond linearly to the control current for low currents (e.g., approximately 1 amp). In other words, control accuracy is not maintained at low currents because the lowered feedback resistance lowers the open loop gain of the controller. This loss of control accuracy comes with the advantage of an improved phase margin at low currents, and is acceptable as long as the slope of the shutdown profile is not too steep. The particular values of high and low coil currents may vary based on the implementation of the present disclosure. In general, the low coil current can be expressed as a percentage of the high coil current. For example, a low coil current may be approximately 10% of the high coil current. 
     Plots of gain and phase versus frequency (e.g., for a range from 1 Hz to 1 MHz plotted according to a log scale) at a high coil current (12 amps (A)) are shown in  FIG. 12 . The curves apply for both the fixed feedback resistance and the variable feedback resistance cases because for a high coil current (e.g., 12 A) they can be the same. As shown, the phase margin (i.e., the difference between the gain and phase curves at zero dB gain) for both the fixed and variable feedback resistance cases at 12 A is approximately 50 degrees (e.g., 54.6 degrees). 
       FIG. 13  illustrates the fixed resistance and variable resistance cases at a low coil current (e.g., 1 A). As can be seen (dotted line), the phase margin for the fixed resistance case has decreased from approximately 50 degrees at 12 A (see  FIG. 12 ) to about 25 degrees (e.g., 24.8 degrees) at 1 A. The reduction in phase margin is due, in part, to a broadening of the bandwidth as the coil current is reduced. The boarding of the bandwidth may be due to operating characteristics of elements in the circuit, such as the IBGT and the inductor (i.e., coil). The phase margin for the variable resistance case, however, has not been reduced nearly as much. The phase margin for the variable resistance case has decreased from approximately 50 degrees at 12 A (see  FIG. 12 ) to about 44 degrees (e.g., 44.3 degrees) at 1 A. Thus, the circuit has a good phase margin for all coil currents during the soft shutdown. Viewed another way, the reduced gain, increases the phase margin at 1 A from about 25 degrees to about 44 degrees. 
     As can be observed from  FIGS. 12 and 13 , the DC gain of circuit may change depending on the coil current and the feedback resistance. For example, a high DC gain at a high current (e.g., 20.7 decibels (dB) at 12 A) may be reduced to a lower DC gain at a low current (e.g., 14.2 dB at 1 A). By changing the feedback resistance the DC gain may be reduced further at a low current (e.g., 5.6 dB at 1 A). The results, shown in  FIGS. 12 and 13  are presented as an example to aid in understanding. In practice, the particular values of coil current, phase, gain, and phase margin may vary. 
     Decreasing the loop gain has advantages over other methods for securing phase margin. For example, internal phase compensation is not appropriate for securing phase margin doe to the low frequencies (a few KHz) required for ignition. In this low frequency regime very large resistors and/or capacitors would be required for the compensation, which are not practical. 
     So far, the discussion of the variable feedback resistor for a negative feedback control loop to provide but accuracy with stability at different current levels has been discuss for applications related to ignition system. In general, however, the principles, techniques, and circuits disclosed herein may be applied to other applications that require a negative feedback loop to limit a current over a range. It is envisioned that the stability and accuracy provided by the variable feedback resistance current limiter circuit is suitable for all such applications. Even more generally, the principles of the present disclosure may be applied to a circuit in which a loop gain is controllable over operating conditions to adjust the accuracy and stability of the output. A schematic of this circuit, in which a loop gain is controllable to adjust accuracy and stability of the output, is shown in  FIG. 14 . 
     In the specification and/or figures, typical embodiments have been disclosed. The present disclosure is not limited to such exemplary embodiments. The use of the term “and/or” includes any and all combinations of one or more of the associated listed items. The figures are schematic representations and so are not necessarily drawn to scale. Unless otherwise noted, specific terms have been used in a generic and descriptive sense and not for purposes of limitation. 
     Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art. Methods and materials similar or equivalent to those described herein can be used in the practice or testing of the present disclosure. As used in the specification, and in the appended claims, the singular forms “a,” “an,” “the” include plural referents unless the context clearly dictates otherwise. The term “comprising” and variations thereof as used herein is used synonymously with the term “including” and variations thereof and are open, non-limiting terms. The terms “optional” or “optionally” used herein mean that the subsequently described feature, event or circumstance may or may not occur, and that the description includes instances where said feature, event or circumstance occurs and instances where it does not. Ranges may be expressed herein as from “about” one particular value, and/or to “about” another particular value. When such a range is expressed, an aspect includes from the one particular value and/or to the other particular value. Similarly, when values are expressed as approximations, by use of the antecedent “about,” it will be understood that the particular value forms another aspect. It will be further understood that the endpoints of each of the ranges are significant both in relation to the other endpoint, and independently of the other endpoint.