Patent Publication Number: US-7902917-B2

Title: Current-input current-output reconfigurable passive reconstruction filter

Description:
BACKGROUND 
     1. Field of the Invention 
     The present invention relates generally to reconstruction filtering. 
     2. Background Art 
     In practice, the output of a digital-to-analog converter (DAC) needs to be band-limited to prevent aliasing. For this reason, a low-pass filter, called reconstruction filter, is typically used at the output of the DAC in order to reduce unwanted images of the information signal. 
     Conventional reconstruction filtering solutions have a number of drawbacks, including being inherently non-linear, using many active elements, and being difficult to reconfigure. These drawbacks make conventional solutions unsuitable for use in multi-band multi-mode wireless transmitters, for example, which require reconstruction filters with high bandwidth programmability and reconfigurability. In addition, conventional solutions are high in cost, power consumption, and area requirements. 
     Accordingly, there is a need for improved reconstruction filtering solutions. 
     BRIEF SUMMARY 
     Embodiments of the present invention relate generally to reconstruction filtering. In particular, embodiments enable highly linear, highly programmable, and easily reconfigurable reconstruction filters. Further, embodiments provide substantial power consumption, area, and cost savings compared to conventional solutions. For example, embodiments use all-passive filtering and substantially reduce active elements compared to conventional solutions. As a result, significant reductions in required area, noise, and power consumption can be achieved. In addition, embodiments perform filtering solely in the current domain, thereby eliminating the non-linear voltage-to-current conversion used in conventional circuits and enabling highly linear filtering. Furthermore, embodiments are highly programmable and easily reconfigurable without the use of tunable capacitors. As such, embodiments are very suitable solutions for multi-band multi-mode wireless transmitters. 
     Further embodiments, features, and advantages of the present invention, as well as the structure and operation of the various embodiments of the present invention, are described in detail below with reference to the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES 
       The accompanying drawings, which are incorporated herein and form a part of the specification, illustrate the present invention and, together with the description, further serve to explain the principles of the invention and to enable a person skilled in the pertinent art to make and use the invention. 
         FIG. 1  illustrates an example wireless transmitter chain. 
         FIG. 2  illustrates a conventional wireless transmitter chain. 
         FIG. 3  illustrates an example wireless transmitter chain according to an embodiment of the present invention. 
         FIG. 4  illustrates an example implementation of a reconstruction filter according to an embodiment of the present invention. 
         FIG. 5  illustrates an example implementation of a reconstruction filter according to an embodiment of the present invention. 
         FIG. 6  illustrates an example implementation of a reconstruction filter according to an embodiment of the present invention. 
         FIG. 7  illustrates an example implementation of a reconstruction filter according to an embodiment of the present invention. 
     
    
    
     The present invention will be described with reference to the accompanying drawings. Generally, the drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number. 
     DETAILED DESCRIPTION OF EMBODIMENTS 
       FIG. 1  illustrates an example wireless transmitter chain  100 . Example transmitter chain  100  may be used in a multi-band multi-mode transmitter. As shown in  FIG. 1 , example transmitter chain  100  includes a digital-to-analog converter (DAC)  102 , a reconstruction filter  104 , an IQ mixer  106 , and a plurality of transmit paths  110   a - d.    
     DAC  102  receives a digital information signal from a baseband processor or a digital module of the transmitter. DAC  102  may be, for example, a 10-bit DAC. DAC  102  converts the received digital information signal from digital to analog. However, because of quantization noise inside DAC  102 , the output of DAC  102  often includes, in addition to a desired baseband information component, undesired images of the baseband information component. The undesired images are typically located at multiples of the sampling frequency of DAC  102 . Accordingly, a reconstruction filter, such as reconstruction filter  104 , is generally used at the output of DAC  102  in order to generate an image-free baseband information signal. 
     The baseband output signal of reconstruction filter  104  is input into an IQ mixer  106 . IQ mixer  106  up-converts the baseband signal to radio frequency (RF). Then, depending on the communication application being used by the transmitter (i.e., frequency band and transmission mode combination), the output of mixer  106  is coupled, via transformers  108   a  and  108   b , to one of the plurality of transmit paths  110 . For example, transmitter chain  100  includes four transmit paths  110   a - d , with transmit paths  110   a - b  configured for high-band applications and transmit paths  110   c - d  configured for low-band applications. Further, within high-band transmit paths  110   a - b  (low-band transmit paths  110   c - d ), transmit path  110   a  ( 110   c ) may be configured for WCDMA signal transmission and transmit path  110   b  ( 110   d ) may be configured for EDGE/GSM signal transmission. Generally, transmit paths  110   a - d  each includes a power amplifier (PA) for amplifying and transmitting the RF signal. 
       FIG. 2  illustrates a conventional wireless transmitter chain  200 . In particular,  FIG. 2  illustrates conventional reconstruction filtering solutions used in wireless transmitter chains. As shown in  FIG. 2 , transmitter chain  200  includes a DAC  102 , a conventional reconstruction filter  202 , and a mixer  106 . 
     Reconstruction filter  202  sits at the output of DAC  102  in order to remove undesired images from the DAC  102  output. Conceptually, as shown in  FIG. 2 , reconstruction filter  202  includes a current-to-voltage conversion stage  204 , a voltage-input voltage-output low pass filtering stage  206 , and a voltage-to-current conversion stage  208 . In practice, this is because DAC  102  initially generates a current output (generally, DAC  102  includes a plurality of current cells), conventional reconstruction filtering is performed in the voltage domain, and mixer  106  requires a current input. 
     Generally, stage  204  includes a trans-impedance circuit that converts a current output signal of DAC  102  to a voltage signal. Stage  206  includes a low-pass filter that filters the voltage output signal of stage  204  and generates a filtered voltage output signal. Stage  208  includes a voltage-to-current converter that converts the voltage output of stage  206  to a current signal. 
     One major drawback of reconstruction filter  202  relates to the inherent non-linearity of voltage-to-current conversion stage  208 . To reduce the non-linear effects of stage  208 , conventional solutions generally include a feedback mechanism implemented using one or more operational amplifiers. In addition, to support complex waveforms such as WCDMA, for example, the filtering stage  206  of reconstruction filter  202  needs to enable at least 3 rd  order filtering. In conventional solutions, this is also implemented using at least three additional operational amplifiers. As a result, four or more operational amplifiers are typically used in conventional reconstruction filters, which results in not only high noise but also very large power consumption and area. 
     Furthermore, to enable multi-mode transmission (e.g., WCDMA, GSM, EDGE, etc.) by the transmitter, the reconstruction filter needs to be highly programmable. For example, for EDGE applications, the required filter bandwidth is only about 100 KHz. For WCDMA applications, on the other hand, the required bandwidth is approximately 2 MHz (20 times the EDGE required bandwidth). Conventional solutions suffer to meet such high programmability requirement primarily because of the large input capacitance of mixer  106 , which acts as a loading capacitance for the reconstruction filter. In general, conventional solutions tend to use tunable capacitors, which are both large and expensive, in order to absorb the loading caused by the input capacitance of the mixer. 
     The above discussed drawbacks of conventional reconstruction filtering call for improved reconstruction filtering solutions. In particular, there is a need for reconstruction filters with small area, low power consumption, and low cost. In addition, reconstruction filters that are highly linear and configurable to support multi-band multi-mode transmitters are needed. 
     Embodiments of the present invention, as will be further described below, satisfy the above desired properties of reconstruction filters. For example, embodiments use all-passive filtering. As a result, active elements, such as operational amplifiers, which introduce high noise and power consumption are eliminated. In addition, embodiments perform filtering solely in the current domain, thereby eliminating the non-linear voltage-to-current conversion used in conventional solutions and enabling highly linear filtering. Furthermore, embodiments are highly programmable and easily reconfigurable without the use of tunable capacitors. As such, embodiments are highly suitable solutions for multi-band multi-mode wireless transmitters. 
       FIG. 3  illustrates an example wireless transmitter chain  300  according to an embodiment of the present invention. For ease of description, only the in-phase (I) branch of transmitter chain  300  is shown in  FIG. 3  and described herein, with the quadrature-phase (Q) branch being identical to the I branch. 
     Example wireless transmitter chain  300  uses a reconstruction filter  302  according to an embodiment of the present invention. It is noted that reconstruction filter  302  represents a conceptual illustration of reconstruction filters according to embodiments of the present invention. 
     As shown in  FIG. 3 , reconstruction filter  302  couples DAC  102  to mixer  106 . In particular, reconstruction filter  302  receives one or more current signals from DAC  102 , and low-pass filters the one or more currents to generate one or more filtered current signals in mixer  106 . In an embodiment, the one or more currents from DAC  102  include a desired information component and undesired image components located at multiples of a sampling frequency of DAC  102 . The one or more filtered current signals contain only the desired information component, and also represent a filtered baseband information signal. It is noted that reconstruction filter  302  filters the DAC currents solely in the current domain to generate the one or more filtered current signals. As such, reconstruction filter  302  represents a current-input current-output filter. Further, reconstruction filter  302  uses no current-to-voltage conversion, voltage-to-voltage filtering, or voltage-to-current conversion as in conventional solutions. As a result, reconstruction filter  302  exhibits very high linearity. 
     In a differential implementation, as shown in  FIG. 3 , reconstruction filter  302  receives a differential current output signal, including currents i+ and i−, from DAC  102 . Accordingly, reconstruction filter  302  includes first and second filtering branches, configured to filter respectively the i+ and i− currents of DAC  102 . In a single-ended implementation, reconstruction filter  302  includes a single filtering branch and receives a single current output signal from DAC  102 . 
     In an embodiment, the first and second filtering branches of reconstruction filter  302  each includes an input stage  304  and a low-pass filter (LPF) stage  306 . Input stages  304   a - b  include one or more input transistors and are used to receive current outputs of DAC  102 . For example, in the embodiment of  FIG. 3 , currents i+ and i− generated by DAC  102  are mirrored respectively into input transistors  304   a  and  304   b  of the first and second filtering branches. 
     LPF stages  306   a - b  provide low-pass filtering of the received DAC currents. In an embodiment, each of LPF stages  306   a - b  includes an all-passive reconfigurable resistor-capacitor (RC) network. The filtered current signals produced by reconstruction filter  302  are generated directly in mixer  106 . In an embodiment, the filtered current output signals of LPF stages  306   a - b  are generated respectively in transistors  308   a - b  of mixer  106 . The filtered current output signals are then mixed with in-phase differential local oscillator (LO) signals, LO, I+ and LO, I−, to generate an up-converted RF signal. 
     According to embodiments of the present invention, the reconfigurable RC network of LPF stages  306   a - b  enables reconstruction filter  302  to have a highly programmable transfer function. For example, on one hand, for simple waveforms reconstruction filter  302  can be programmed to operate as a single order filter. On the other hand, for complex waveforms, such as WCDMA, for example, reconstruction filter  302  can be programmed to operate as a third-order filter (e.g., third-order Butterworth). 
     In addition, the RC network of LPF stages  306   a - b  enables reconstruction filter  302  to be easily reconfigurable. In an embodiment, the transfer function of reconstruction filter  302  can be reconfigured by varying a time constant of the RC network. In another embodiment, the transfer function of reconstruction filter  302  can be reconfigured by simply varying one or more resistors in the RC network. 
     Because LPF stages  306   a - b  use an all passive RC network, reconstruction filter  302  has a minimum of active elements (e.g., operational amplifiers), which significantly reduces power consumption, area, and cost requirements of embodiments of the present invention compared to conventional solutions. In addition, in embodiments, only device capacitors (e.g., MOS capacitors) are used in the RC network. As such, the loading caused by the large input capacitance of mixer  106  can be absorbed, without affecting the programmability of reconstruction filter  102 . This is in addition to requiring less area by using device capacitors. 
     Example implementations of reconstruction filter  302  according to embodiments of the present invention are described below with reference to  FIGS. 4-7 . As would be understood by a person skilled in the art based on the teachings herein, embodiments of the present invention are not limited to the example implementations provided herein, but extend to any variations and or improvements which would be apparent to a person skilled in the art. 
       FIG. 4  illustrates an example implementation  400  of a reconstruction filter according to an embodiment of the present invention. In particular,  FIG. 4  shows a reconstruction filter according to an embodiment of the present invention coupled to the input stage of a subsequent circuit. In an embodiment, the subsequent circuit includes a mixer, as shown above in  FIG. 3 . 
     According to example implementation  400 , the reconstruction filter includes an input node  402 , an input transistor  406 , and an all-passive resistor-capacitor (RC) network  408 . The reconstruction filter is coupled to an input stage of a mixer, which includes an input capacitor C 2    418  and an input transistor  420 . Generally, input capacitor C 2    418  has a very large capacitance. 
     Input node  402  receives an input current signal i S    404 . In an embodiment, input current signal i S    404  is received from a digital-to-analog converter (DAC) and contains a desired information component and undesired image components located at multiples of a sampling frequency of the DAC. 
     Input transistor  406  has a first terminal coupled to input node  402  and a second terminal coupled to ground. A third terminal of input transistor  406  is coupled to RC network  408 , which is also coupled to the input stage of the mixer. In an embodiment, input transistor  406  forms a current mirror circuit with input transistor  420  of the mixer coupled to the reconstruction filter. 
     In an embodiment, RC network  408  includes a first capacitor C 3    410  having a first end coupled to the third terminal of the input transistor  406  and a second end coupled to ground; a first resistor R 1    412  having a first end coupled to the first end of first capacitor C 3    410  and a second end coupled to input node  402 ; a second capacitor C 1    414  having a first end coupled to input node  402  and a second end coupled to ground; and a second resistor R 2    416  having a first end coupled to the first end of first resistor R 1    412  and a second end coupled to the subsequent circuit coupled to the reconstruction filter circuit. In an embodiment, the second end of R 2    416  is coupled to a gate terminal of input transistor  420  of the mixer. 
     In an embodiment, first capacitor C 3    410  and second capacitor C 1    414  are matched to input capacitor C 2    418  of the mixer. This reduces variations in output current signal i O    422  that are due to temperature/process variations in the first and second capacitors and the input capacitor of the mixer. In another embodiment, first capacitor C 3    410  and second capacitor C 1    414  have substantially equal DC bias as input capacitor C 2    418  of the mixer. In addition, in embodiments, only device capacitors (e.g., MOS capacitors) are used in RC network  408 . As such, the loading caused by the large capacitance of input capacitor C 2    418  can be absorbed, without affecting the programmability of reconstruction filter  102 . This is in addition to requiring less area by using device capacitors. 
     In an embodiment, the reconstruction filter filters input current signal i S    404  to generate an output current signal i O    422  in input transistor  420  of the mixer. Output current signal i O    422  contains only the desired information component of input current signal i S    404 . In an embodiment, output current signal i O    422  represents a filtered baseband information signal, which is subsequently upconverted by the mixer to RF. 
     Note that according to example implementation  400 , the reconstruction filter includes a feedback loop, formed by the coupling of the second end of first resistor R 1    412  to input node  402 . As such, the reconstruction filter can have complex poles and may thus be configured as a Butterworth filter, for example. In an embodiment, the reconstruction filter is programmable to operate as a third-order Butterworth filter. 
     In an embodiment, a transfer function of the reconstruction filter of  FIG. 4  can be written as: 
               H   ⁡     (   s   )       =         V   O       i   s       =     1                   R   1     ⁢     R   2     ⁢     C   1     ⁢     C   2     ⁢     C   3     ⁢     s   3       +       (         R   2     ⁢     C   2     ⁢     C   3       +       R   1     ⁢     C   1     ⁢     C   2       +       R   1     ⁢     C   1     ⁢     C   3       +       R   2     ⁢     C   1     ⁢     C   2         )     ⁢     s   2       +                       ⁢         (       C   1     +     C   2     +     C   3     +       g   m     ⁢     R   2     ⁢     C   2         )     ⁢   s     +     g   m                         
where V O  represents the voltage at the input of the mixer, as shown in  FIG. 4 .
 
     According to embodiments, the transfer function of the reconstruction filter is programmable by varying a time constant of RC network  408 . In an embodiment, the transfer function of the reconstruction filter is programmable solely by varying one or more of first resistor R 1    412  and second resistor R 2    416 . In an embodiment, first resistor R 1    412  and second resistor R 2    416  are implemented as switch-programmable resistors, which enables the reconstruction filter to be easily reconfigurable. On the other hand, first capacitor C 3    410  and second capacitor C 1    414  are implemented as fixed value capacitors. 
     As described above, the reconstruction filter filters input current signal i S    404  solely in the current domain to generate output current signal i O    422 . As such, embodiments avoid current-to-voltage and voltage-to-current conversion used in conventional solutions, and enable highly linear filtering. In addition, it is noted that any noise caused by input transistor  406  is filtered in the same manner as input current signal i S    404 . Accordingly, output current signal i O    422  exhibits very low noise levels. 
       FIG. 5  illustrates another example implementation  500  of a reconstruction filter according to an embodiment of the present invention. Similar to  FIG. 4 ,  FIG. 5  shows a reconstruction filter according to an embodiment of the present invention coupled to the input stage of a subsequent circuit. In an embodiment, the subsequent circuit includes a mixer, as shown above in  FIG. 3 . 
     Example implementation  500  is structurally similar to example implementation  400 , but also includes an output transistor  502 , as shown in  FIG. 5 . Output transistor  502  has a drain terminal coupled to input node  402 , a source terminal coupled to ground, and a gate terminal coupled to the gate terminal of input transistor  420  of the mixer. 
     The transfer function of the reconstruction filter of  FIG. 5  can be written as: 
                     ⁢       H   ⁡     (   s   )       =         V   O       i   s       =           ⁢     1                   R   1     ⁢     R   2     ⁢     C   1     ⁢     C   2     ⁢     C   3     ⁢     s   3       +       (         R   2     ⁢     C   2     ⁢     C   3       +       R   1     ⁢     C   1     ⁢     C   2       +       R   1     ⁢     C   1     ⁢     C   3       +       R   2     ⁢     C   1     ⁢     C   2         )     ⁢     s   2       +                       ⁢         (       C   1     +     C   2     +     C   3     +       g     m   ⁢           ⁢   1       ⁢     R   2     ⁢     C   2         )     ⁢   s     +     g     m   ⁢           ⁢   1       +     g     m   ⁢           ⁢   2                             
where V O  represents the voltage at the input of the mixer, as shown in  FIG. 5 .
 
     In an embodiment, input transistor  406  and output transistor  502  are matched according to a transconductance ratio, which allows example implementation  500  to have a smaller area compared to example implementation  400 . Further, example implementation  500  may be suited for cases in which input current signal i S    404  has a small AC signal swing (e.g., less than 25%). 
       FIG. 6  illustrates another example implementation  600  of a reconstruction filter according to an embodiment of the present invention. Similar to  FIG. 4 ,  FIG. 6  shows a reconstruction filter according to an embodiment of the present invention coupled to the input stage of a subsequent circuit. In an embodiment, the subsequent circuit includes a mixer, as shown above in  FIG. 3 . 
     Example implementation  600  is structurally similar to example implementation  400 , but also includes a diode-connected transistor  602  having a first terminal and a second terminal coupled to input node  402  and a third terminal coupled to ground. The transfer function of the reconstruction filter of  FIG. 6  can be written as: 
               H   ⁡     (   s   )       =           ⁢         V   O       i   s       =     1                     R   1     ⁢     R   2     ⁢     C   1     ⁢     C   2     ⁢           ⁢     C   3     ⁢           ⁢     s   3       +           ⁢     (         R   2     ⁢     C   2     ⁢     C   3       +           ⁢       R   1     ⁢     C   1     ⁢     C   2       +       R   1     ⁢     C   1     ⁢     C   3       +       R   2     ⁢     C   1     ⁢     C   2       +                           g     m   ⁢           ⁢   2       ⁢     R   1     ⁢     R   2     ⁢     C   2     ⁢     C   3       )     ⁢     s   2       +           ⁢     (       C   1     +     C   2     +     C   3     +       g     m   ⁢           ⁢   1       ⁢     R   2     ⁢     C   2       +                           g     m   ⁢           ⁢   2       ⁡     (         R   1     ⁢     C   3       +       R   2     ⁢     C   2       +       R   1     ⁢     C   2         )       )     ⁢   s     +     g     m   ⁢           ⁢   1       +     g     m   ⁢           ⁢   2                           
where V O  represents the voltage at the input of the mixer, as shown in  FIG. 6 .
 
       FIG. 7  illustrates another example implementation  700  of a reconstruction filter according to an embodiment of the present invention. Similar to  FIG. 4 ,  FIG. 7  shows a reconstruction filter according to an embodiment of the present invention coupled to the input stage of a subsequent circuit. In an embodiment, the subsequent circuit includes a mixer, as shown above in  FIG. 3 . 
     Example implementation  700  is structurally similar to example implementation  400 , but also includes an operational amplifier  702  coupled between input transistor  406  and RC network  408 . As such, the first and third terminals of input transistor  406  are coupled to one another and also to the inverting input of operational amplifier  702 . The non-inverting input of operation amplifier  702  is coupled to the second end of first resistor R 1    412 , forming a feedback loop in the reconstruction filter. The output terminal of operational amplifier  702  is coupled to the first end of first resistor R 1    412 . 
     The transfer function of the reconstruction filter of  FIG. 7  can be written as: 
     
       
         
           
             
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     As would be understood by a person skilled in the art based on the teachings herein, example implementations  500 ,  600 , and  700  have similar functionality, operation, and performance as described above with respect to example implementation  400 . In addition, similar implementations as described above with respect to example implementation  400  can be applied in example implementations  500 ,  600 , and  700 . 
     It is to be appreciated that the Detailed Description section, and not the Summary and Abstract sections, is intended to be used to interpret the claims. The Summary and Abstract sections may set forth one or more but not all exemplary embodiments of the present invention as contemplated by the inventor(s), and thus, are not intended to limit the present invention and the appended claims in any way. 
     Embodiments of the present invention has been described above with the aid of functional building blocks illustrating the implementation of specified functions and relationships thereof. The boundaries of these functional building blocks have been arbitrarily defined herein for the convenience of the description. Alternate boundaries can be defined so long as the specified functions and relationships thereof are appropriately performed. 
     The foregoing description of the specific embodiments will so fully reveal the general nature of the invention that others can, by applying knowledge within the skill of the art, readily modify and/or adapt for various applications such specific embodiments, without undue experimentation, without departing from the general concept of the present invention. Therefore, such adaptations and modifications are intended to be within the meaning and range of equivalents of the disclosed embodiments, based on the teaching and guidance presented herein. It is to be understood that the phraseology or terminology herein is for the purpose of description and not of limitation, such that the terminology or phraseology of the present specification is to be interpreted by the skilled artisan in light of the teachings and guidance. 
     The breadth and scope of embodiments of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.