Patent Publication Number: US-2023155692-A1

Title: Method and device for energy transfer and harvesting

Description:
FIELD AND BACKGROUND 
     This disclosure invention relates to a method and device for energy transfer and/or harvesting, more particularly but not exclusively, for powering wearable or implantable electronics using energy transmission or harvesting via the human body. 
     Despite advances in ultra-low-power circuit techniques and battery energy density, sustaining mm-sized chips of sub-mW power consumption purely using a battery would still require periodic replacement/recharging, which not only causes interruptions and inconveniences but also affects functionality especially for wearable electronic devices such as earbuds, smart band-aids, and electrocardiography (ECG) sensors to name but a few. 
     Near-field power transmission approaches using an inductive link impose stringent requirements on alignment and are typically designed for short distances. RF-based power transmission covers a longer distance. However, when used in the body area, it experiences ˜20 dB path loss degradation with the antenna pattern distorted, and a further 20-40 dB reduction in the path gain under non-line-of sight (non-LoS) transmission, where the human body blocks the propagation path. 
     A more sustainable alternative is to replace or at least complement battery by continually extracting energy from the ambient environment, or through human-environment interactions. Commonly used power harvesting sources for wearable devices include photovoltaics, thermo-, piezo-, and triboelectricity, all of which require either solar panels (for photovoltaics) or nanogenerators for power extraction, introducing 1) inherent area cost at each node and 2) limitation of placement due to the way in which they operate. 
     Additionally, as harvestable power decreases almost proportionally with harvester downsizing, a tension between power and size becomes inevitable. Moreover, these potential power sources are not equally distributed around the human body, leaving some sensor nodes unsustainable. One example concerns piezo- and triboelectricity harvesting. While 10-100 μW could generally be obtained in locations where constant movements take place such as joints or heels, it is expected that a sharp decrease would occur for devices placed on chest or head, a conventional placement in a body area network. Apart from the harvester location dependency, the amount of power extractable is also prone to clothing and environmental changes. 
     Electromagnetic (EM) waves, on the other hand, are omnipresent in modern life. From natural and artificial lights to the cellular network, to power cables and electronic gadgets, EM fields of various frequency bands have always been all around. RF band EM power harvesting has attracted growing attention, and the scavenging of 2.4 GHz WiFi radio transmitted in the air as low as −36 dBm has been demonstrated as viable. However, unlike the case of RF power harvesting/delivery in the air, RF power transfer to body area electronic nodes suffers from severe performance degradation and is thus unsuitable for wearable applications. First of all, RF-based harvesting/transfer is vulnerable to the body-shadowing effect, where the human body absorbs the majority of RF energy. Thus, RF harvesting around the body requires a line-of-sight (LOS) condition between the source and the receiver, which is challenging to guarantee under a body area network (BAN) application scenario. 
     Moreover, an antenna placed in vicinity of the body undergoes significant changes in radiation patterns as compared to when in the air, mainly due to the distinct dielectric properties of the human skin surface. Antenna gain also decreases by around 20 dB, even under the loss scenario. For RF-based ambient harvesting, in particular, that is a significant degradation considering the incident power of generally less than −20 dBm available to the receiving antenna. 
     Apart from the RF waves present primarily due to wireless communication, ambient fields lying towards the lower end of the EM spectrum are just as ubiquitous and abundant. Sources for these fields include AM radio signals, underground telecommunication cables, and most dominantly, electricity networks together with appliances or utilities linked in for power supply. However, as antenna size is inversely proportional to wave frequency, harvesting at super-low-frequencies such as 50/60 Hz would necessitate an antenna size of 3000 km, as compared with around 6.25 cm for RF signals. 
     It is desirable to provide a method and device for transmitting and/or harvesting energy which addresses at least one of the drawbacks of the prior art and/or to provide the public with a useful choice. 
     SUMMARY 
     In a first aspect, there is provided a power receiver comprising: a first electrode arranged to be electrically coupled to a body of a living being, the first electrode operable to receive an electrical signal via the body; and a rectifier for converting the received electrical signal into a rectified electrical signal, the rectifier having a plurality of rectifier switches and operable in a bulk biasing mode in which selected rectifier switches of the plurality of rectifier switches are forward bulk biased. 
     As described in the preferred embodiment, forward bulk biasing certain switches of the rectifier depending on voltage may improve the performance of the rectifier at low input voltage and therefore increase the sensitivity of the receiver and enable power transfer over long distances and power harvesting. 
     It should be appreciated that the use of the term ‘electrode’ is meant to include any electrical contact including electrical connectors of integrated circuits, or suitable electrical contacts which may be implantable within a living being body particular if the power receiver is an implantable device. 
     Advantageously, the rectifier may be operable selectively between the bulk biasing mode and a non-bulk biasing mode in which the selected rectifier switches are not forward bulk biased, in dependence on an amplitude of the rectified electrical signal. 
     In a specific embodiment, second selected rectifier switches of the plurality of rectifier switches may be reverse bulk biased in the bulk biasing mode, and not reversed bulk biased in the non-bulk biased mode which may help to prevent current leakage. The selected rectifier switches and the second selected rectifier switches may alternately forward and reversed bulk biased in the bulk biasing mode, in dependence on a phase of the electrical signal. The rectifier may be operable in the bulk biasing mode at amplitudes of the rectified electrical signal less than or equal to 0.4V and in the non-bulk biased mode otherwise. 
     The power receiver may have an input load impedance and further comprise an L-C circuit, the input load impedance being dependent on a resonance response of the L-C circuit. The L-C circuit may comprise a parallel L-C circuit. This may help to improve the power received at the receiver. 
     The electrical signal may have a power level and the power receiver may further comprise a DC-DC converter for converting the rectified electrical signal to an output electrical signal, the DC-DC converter having a number of selectable discontinuous mode (DCM) times and operable between a power transmission operation mode in which the electrical signal is transmitted by a power transmitter; and a power harvesting operation mode in which the electrical signal is harvested ambient energy, in dependence on the power level of the electrical signal. The power receiver may also comprise a controller arranged to select a discontinuous mode (DCM) time from the number of selectable discontinuous mode (DCM) times for the DC-DC converter in response to the operation mode of the DC-DC converter. Switching between power receiving modes and power harvesting modes by altering the discontinuous mode (DCM) time may facilitate switching between modes and enable optimal power recovery in both modes. The controller may be arranged to select between a plurality of power transmission discontinuous mode (DCM) times in response to the power transmission operation mode, and between a plurality of power harvesting discontinuous mode (DCM) times in response to the power harvesting operation mode, in dependence of a voltage of the rectified electrical signal, thereby facilitating optimal power transmission according to the voltage of the signal. 
     A switching frequency of the DC-DC converter in the power transmission operation mode may be in the range 10 kHz to 500 kHz and a switching frequency of the DC-DC converter in the power harvesting operation mode may be in the range 500 Hz to 6 kHz. 
     The controller may be further arranged to adjust an inductor charging time of the DC-DC converter within a predetermined range until an input power of the DC-DC converter reaches a maximum value for the predetermined range. The controller may be arranged to determine a direction of adjustment of the inductor charging time based on a corresponding inductor discharging time of the DC-DC converter. The inductor charging time of the DC-DC converter may be between 100 ns and 500 ns. Adjustment of the inductor charging time in this way may enable straightforward fine tuning of the switching frequency of the converter to optimize power recovery. 
     The power receiver may be formed as an integrated circuit (IC) chip. 
     The power receiver may further comprise a floating ground node operable as a parasitic capacitive return path via an external ground for the electrical signal, and the first electrode may be operable to receive the electrical signal via the body through capacitive coupling. The first electrode may instead comprise a receiving coil, and the first electrode may be operable to receive the electrical signal via the body through magnetic resonance coupling with a transmitting coil. The power receiver may instead further comprise a second electrode arranged to be electrically coupled to the body of the living being, and wherein the electrical signal is galvanic current. In a second aspect, there is provided a power receiver comprising: a first electrode arranged to be electrically coupled to a body of a living being, the first electrode operable to receive an electrical signal via the body, the electrical signal having a power level; a rectifier for rectifying the electrical signal into a rectified electrical signal; a DC-DC converter for converting the rectified electrical signal to an output electrical signal, the DC-DC converter having a number of selectable discontinuous mode (DCM) times and operable between a power transmission operation mode in which the electrical signal is transmitted by a power transmitter; and a power harvesting operation mode in which the electrical signal is harvested ambient energy, in dependence on the power level of the electrical signal; and a controller arranged to select a discontinuous mode (DCM) time from the number of selectable discontinuous mode (DCM) times for the DC-DC converter in response to the operation mode of the DC-DC converter. 
     As described in the preferred embodiment, switching between power receiving modes and power harvesting modes by altering the discontinuous mode (DCM) time may facilitate switching between modes and enable optimal power recovery in both modes. 
     In a third aspect, there is provided a power transmitter comprising: a signal generator having a variable output impedance; the signal generator operable to generate an electrical signal; a first electrode arranged to be electrically coupled to a living being body for transmission of the electrical signal via the living being body; an amplitude detector configured to measure an amplitude of the electrical signal; and a controller operable to adjust an impedance of the variable output impedance based on the amplitude of the electrical signal to enable the amplitude of the electrical signal to at least reach a threshold amplitude for transmission by the first electrode. 
     As described in the preferred embodiment, adjusting the output impedance based on signal amplitude may enable the transmitter to perform impedance matching of the load due to both the human body and the environment and also detect changes in the load and respond accordingly, which may lead to improved power transmission over long distances. With transmitters according to this embodiment, impedance monitoring, including at the TX-interface, may be performed without introducing design complexity and a reduction in the reflection coefficient may be achieved. 
     The controller may be operable to determine that the electrical signal has at least reached the threshold amplitude when a magnitude of a difference between the amplitude of the electrical signal and a half open circuit amplitude of the transmitter is less than a predetermined value. 
     The variable output impedance may comprise a resistive component and a capacitive component, and the controller may be operable to adjust the impedance of the variable output impedance based on a multichotomic search of the amplitude of the electrical signal within a two-dimensional space described by the capacitive component and the resistive component of the variable output impedance to enable the amplitude of the electrical signal to at least reach the threshold amplitude for transmission by the first electrode. The multichotomic search of the amplitude of the electrical signal within two-dimensional space described by the capacitive component and the resistive component of the variable environment output impedance may comprise dividing the two-dimensional space into a plurality of areas and determining the amplitude of the electrical signal at a single point in each of the plurality of areas; and selecting the area for which the amplitude of the electrical signal at the single point is the highest closest to the half the open circuit output amplitude of the transmitter to be searched in a following stage of the multichotomic search. 
     The signal generator may comprise a first capacitor bank having a first variable configuration, and the capacitive element of the variable output impedance may be dependent on the first configuration of the first capacitor bank; and may further comprise a second capacitor bank having a second variable configuration, in which the resistive component of the variable output impedance may be dependent on the second configuration of the second capacitor bank, wherein the controller being operable to change at least one of the first configuration and the second configuration to adjust the impedance value of the variable output impedance. 
     The transmitter may be formed as an integrated circuit (IC) chip. 
     The power transmitter may further comprise a floating ground node operable as a parasitic capacitive return path via an external ground for the electrical signal, such that the first electrode is operable to transmit the electrical signal via the body through capacitive coupling. The first electrode may instead comprise a transmitting coil, the transmitting coil being operable to transmit the electrical signal via the body through magnetic resonance coupling with a receiving coil. The power receiver may instead further comprise a second electrode arranged to be electrically coupled to the body of the living being, wherein the electrical signal is galvanic current. 
     In a fourth aspect, an energy transfer apparatus may be provided. The energy transfer apparatus may comprise a power transmitter comprising: a signal generator operable to generate an electrical signal, and an electrode arranged to be electrically coupled to a living being body for transmission of the electrical signal via the living being body; and a power receiver comprising: a first electrode arranged to be electrically coupled to a body of a living being, the first electrode operable to receive an electrical signal via the body; a rectifier for converting the received electrical signal into a rectified electrical signal, the rectifier having a plurality of rectifier switches and operable selectively between a bulk biasing mode in which selected rectifier switches of the plurality of rectifier switches are forward bulk biased, and a non-bulk biased mode in which the selected rectifier switches are not forward bulk biased, in dependence on an amplitude of the rectified electrical signal. The signal generator may have an output load impedance, the power transmitter further comprising an L-C circuit, the output load impedance being dependent on a resonance response of the L-C circuit. The power transmitter may instead have a variable output impedance and further comprise: an amplitude detector configured to measure an amplitude of the electrical signal; and a controller operable to adjust an impedance of the variable output impedance based on the amplitude of the electrical signal to enable the amplitude of the electrical signal to at least reach a threshold amplitude for transmission by the electrode. 
     In a fifth aspect, there is provided an energy transfer apparatus which comprises: a signal generator having a variable output impedance; the signal generator operable to generate an electrical signal; a first electrode arranged to be electrically coupled to a living being body for transmission of the electrical signal via the living being body; an amplitude detector configured to measure an amplitude of the electrical signal; and a controller operable to adjust an impedance of the variable output impedance based on the amplitude of the electrical signal to enable the amplitude of the electrical signal to at least reach a threshold amplitude for transmission by the first electrode. The energy transfer apparatus of this aspect further comprises a power receiver comprising: an electrode arranged to be electrically coupled to a body of a living being, the electrode operable to receive an electrical signal via the body, and a rectifier for rectifying the received electrical signal into a rectified electrical signal. The power receiver further may further comprise an L-C circuit, the input load impedance being dependent on a resonance response of the L-C circuit. 
     In a sixth aspect, there is provided a method of receiving electrical power via an electrode coupled to a body of a living being, the method comprising: receiving an electrical signal from the body via the electrode; rectifying the received electrical signal into a rectified electrical signal using a rectifier, the rectifier having a bulk biasing mode in which selected rectifier switches of a plurality of rectifier switches are forward bulk biased. 
     Advantageously, the rectifier may include a non-bulk biased mode in which the selected rectifier switches are not forward bulk biased, and the method may include selectively operating the rectifier between the bulk biasing mode and the non-bulk biasing mode, in dependence on an amplitude of the rectified electrical signal. 
     The method may further comprise varying an input load impedance for receiving the electrical signal in accordance with a resonance response of an L-C circuit. Alternatively, or additionally, the method may further comprise measuring a power level of the electrical signal; converting the rectified electrical signal to an output electrical signal using a DC-DC converter, the DC-DC converter having a number of selectable discontinuous mode (DCM) times and operable between a power transmission operation mode in which the electrical signal is transmitted by a power transmitter; and a power harvesting operation mode in which the electrical signal is harvested ambient energy, in dependence on the power level of the electrical signal; and selecting a discontinuous mode (DCM) time from the number of selectable discontinuous mode (DCM) times for the DC-DC converter in response to the operation mode of the DC-DC converter. 
     The method may further comprise adjusting an inductor charging time of the DC-DC converter within a predetermined range to until an input voltage of the DC-DC converter reaches a maximum value for the predetermined range. The method may further comprise determining an inductor discharging time of the DC-DC converter; determining a direction of adjustment of the inductor charging time, based on the inductor discharging time; and adjusting the inductor charging time of the DC-DC converter in the direction of adjustment. 
     In a seventh aspect, a method of receiving electrical power via an electrode coupled to a body of a living being is provided in which the method comprises: receiving an electrical signal from the body via the electrode; measuring a power level of the electrical signal; rectifying the electrical signal to obtain a rectified electrical signal; converting the rectified electrical signal to an output electrical signal using a DC-DC converter, the DC-DC converter having a number of selectable discontinuous mode (DCM) times and operable between a power transmission operation mode in which the electrical signal is transmitted by a power transmitter; and a power harvesting operation mode in which the electrical signal is harvested ambient energy, in dependence on the power level of the electrical signal; and selecting a discontinuous mode (DCM) time from the number of selectable discontinuous mode (DCM) times for the DC-DC converter in response to the operation mode of the DC-DC converter. 
     In an eighth aspect, there is provided a method of transmitting electrical power via an electrode coupled to a body of a living being, in which the method comprises: generating, by a signal generator having a variable output impedance, an electrical signal; measuring an amplitude of the electrical signal; adjusting an impedance value of the variable output impedance based on the amplitude of the electrical signal to enable the amplitude of the electrical signal to at least reach a threshold amplitude; and transmitting the electrical signal via the electrode to the living being body. The variable output impedance may have a resistive and a capacitive component, and the method may further comprise: performing a multichotomic search of the amplitude of the electrical signal within a two-dimensional space described by the capacitive component and the inductive resistive component of the variable output impedance; and adjusting the impedance value of the variable output impedance based on the multichotomic search. Performing the multichotomic search of the amplitude of the electrical signal may further comprise dividing the two-dimensional space into a plurality of areas and determining the amplitude of the electrical signal at a single point in each of the plurality of areas; and selecting the area for which the amplitude of the electrical signal at the single point is the highest to be searched in a following stage of the multichotomic search. 
     In a ninth aspect, a method of transferring power between a power transmitter and a power receiver is provided. The power transmitter includes a signal generator having an output impedance circuit, and a first transmitter electrode electrically coupled to a living being body; the power receiver including a rectifier and a first receiver electrode electrically coupled to the living being, the method comprising: generating an electrical signal by the signal generator, the signal generator having an output impedance, wherein; an impedance of the output impedance is adjusted based on one of resonance response of an L-C circuit, and an amplitude of the generated electrical signal; transmitting the electrical signal via the first transmitter electrode to the living being body; receiving the transmitted electrical signal, by the first receiver electrode, via the living being body; and rectifying the received electrical signal to produce a rectified electrical signal. 
     The output impedance circuit may further include a floating transmitter node; and the power receiver may further include a floating receiver node; the first transmitter electrode arranged to cooperate with the first receiver electrode to form a transmission path via the living being body for the electrical signal, and the floating transmitter node being arranged to cooperate with the floating receiver node to form a parasitic capacitive return path via an external ground for the electrical signal. 
     Alternatively, the first transmitter electrode may include a transmitting coil and the first receiver electrode may include a receiving coil, wherein the power transmitter is operable to transmit the electrical signal to the power receiver via the living being body through magnetic resonance coupling between the transmitting and the receiving coil. 
     Further alternatively, the power transmitter may further include a second transmitter electrode coupled to the living being body, and the power receiver further may include a second receiver electrode, wherein the electrical signal being transmitted by the first transmitter electrode is galvanic current. 
     The method may further comprise adjusting an input load impedance of the power receiver in dependence on a predetermined resonance response. The rectifier may have a bulk biasing mode in which first selected rectifier switches of a plurality of rectifier switches are forward bulk biased, and a non-bulk biased mode in which the first selected rectifier switches are not forward bulk biased, and rectifying the electrical signal may comprise: selectively operating the rectifier between the bulk biasing mode and the non-bulk biasing mode, in dependence on an amplitude of the rectified electrical signal. The method may further comprise converting the rectified electrical signal to an output electrical signal using a DC-DC converter, the DC-DC converter having a number of selectable discontinuous mode (DCM) times; and selecting a discontinuous mode (DCM) time from the number of selectable discontinuous mode (DCM) times for the DC-DC converter in dependence of a voltage of the rectified electrical signal. Yet further, the method may comprise measuring an amplitude of the electrical signal; and adjusting the output impedance of the signal generator based on the amplitude of the electrical signal to enable the amplitude of the electrical signal to at least reach a threshold amplitude. 
     It should be apparent that features relating to one aspect may also be applicable to the other aspects. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Exemplary embodiments will now be described with reference to the accompanying drawings, in which: 
         FIG.  1    shows an energy transfer apparatus according to a preferred embodiment coupled to a person; 
         FIG.  2   a    shows a schematic representation of the energy transfer apparatus of  FIG.  1    in a power receiving mode; 
         FIG.  2   b    shows the apparatus of  FIGS.  1  and  2     a  in a power harvesting mode; 
         FIG.  3    shows a schematic overview of the transmitter and one of the receivers of the apparatus of  FIG.  2   a    and  FIG.  2     b;    
         FIG.  4    shows a block diagram of the power transmitter of the apparatus of  FIG.  2     a;    
         FIG.  5    shows an impedance range covered by the dynamic impedance matching of the power transmitter of  FIG.  4   ; 
         FIG.  6    shows a method of impedance matching performed by the power transmitter of  FIG.  4   ; 
         FIG.  7    shows a change in output power of the power transmitter of  FIG.  4    as impedance matching according to the method of  FIG.  6    is performed; 
         FIG.  8    shows a detuned impedance booster of the receiver of  FIG.  3   ; 
         FIG.  9    shows an example impedance boosting performed by the detuned impedance booster of  FIG.  8   ; 
         FIG.  10    shows a bulk adaptation rectifier of the receiver of  FIG.  3   ; 
         FIG.  11    illustrates two modes of operation of the bulk adaptation rectifier of  FIG.  10   ; 
         FIG.  12    illustrates source-bulk and bulk-source behaviors of the bulk adaptation rectifier of  FIG.  11   ; 
         FIG.  13   a    shows a simplified architecture of the Dual Mode Buck-Boost Converter (DM-BCC) of the receiver of  FIG.  3   ; 
         FIG.  13   b    shows a schematic representation of the expected PHI 1 , PHI 2  signals and resulting current I L  across the inductor of the switch matrix of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     a;    
         FIG.  14    shows a schematic of the T 1  and T 2  delay modules of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     a;    
         FIG.  15    schematically illustrates the Adaptive level shifter (ALS)  1309  of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     b;    
         FIG.  16    schematically illustrates the T 0  delay module of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     b;    
         FIGS.  17   a  and  17   b    show the effect of the configurations of the T 0  delay module of  FIG.  16    on conversion efficiency in the power transfer and harvesting modes, respectively; 
         FIG.  17   c    shows an outline of the method of the tuning of T 1  and T 2  of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     b;    
         FIG.  17   d    shows a method of adjustment of T 1  performed by the MPPT module of the Dual Mode Buck-Boost Converter (DM-BCC) of  FIG.  13     a;    
         FIG.  18    shows a change of S 11  of a body coupled transmitter with fixed impedance matching according to environment; 
         FIG.  19    shows power recovery as a function of distance with and without dynamic impedance matching as performed by the transmitter of  FIG.  4   ; 
         FIG.  20    shows an effect of the Bulk Adaptation Rectifier of  FIG.  10    on recovered power; 
         FIGS.  21   a ,  21   b  and  21   c    show recovered power as a function of distance for three different receivers, including the receiver according to the preferred embodiment; 
         FIG.  22    shows an effect of impedance load at the receiver on power recovered in both power transmission and harvesting; 
         FIG.  23    shows a power spectrum of the ambient EM waves coupled on the human body and in the air; 
         FIG.  24    shows recovered ambient power in different environments using a receiver according to the preferred embodiment; 
         FIG.  25    shows recovered ambient power at different locations on the body using a receiver according to the preferred embodiment; 
         FIG.  26    shows a power consumption of an example converter as shown in  FIG.  13   a    at different T 0  configurations; 
         FIG.  27    shows a power consumption breakdown of an example converter as shown in  FIG.  13   a    at different T 0  configurations; 
         FIG.  28    shows a variation of the preferred embodiment with a transmitter with a static impedance matching output interface; 
         FIG.  29    shows an energy transfer apparatus according to a second embodiment; 
         FIG.  30    shows experimental results for the recovered power as a function of transmitter output voltage obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  31    shows experimental results for the recovered power as a function of transmitter-receiver distance obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  32    shows experimental results for the recovered power as a function of the number of receivers as obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  33    shows experimental results for the recovered power as a function of electrode size as obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  34    shows wrist-to-wrist channel gain measured on three different subjects as obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  35    shows channel gain along the human arm as obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  36    shows channel gain measured along heterogeneous body path as obtained for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  37    shows the body coupled power received for an RF generator; 
         FIG.  38    shows the body coupled power received for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  39    shows the human body channel gain in comparison with RF approach for apparatus according to the second embodiment of  FIG.  29   ; 
         FIG.  40    shows the recovered power with and without the impedance/voltage boosting techniques using the L-C tank of the second embodiment of  FIG.  29   ; 
         FIG.  41    shows the open circuit voltage with and without the impedance/voltage boosting techniques using the L-C tank of the second embodiment of  FIG.  29   ; 
         FIG.  42    shows the improvement in power recovery at a range of power transmission frequencies using the L-C tank of the second embodiment of  FIG.  29   ; 
         FIG.  43    shows power recovery along an on-body path from one wrist to the other wrist using the L-C tank of the second embodiment of  FIG.  29   ; 
         FIGS.  44  to  46    show the load impedance for maximum power extraction obtained using receivers according to the second embodiment of  FIG.  29   ; 
         FIG.  47    shows power harvested with a 30 MΩ load resistance under seven environmental settings extraction obtained using a receiver according to the second embodiment of  FIG.  29   ; 
         FIG.  48    shows power harvested with a 30 MΩ load resistor by 5 receivers according to the second embodiment of  FIG.  29    placed on different parts of the body simultaneously; 
         FIG.  49    shows power harvested using a receiver according to the second embodiment of  FIG.  29    with a 30 MΩ resistor and the open-circuit voltage achieved with various electrode sizes; 
         FIG.  50    shows power harvested using a receiver according to the second embodiment of  FIG.  29    from 4 different environmental scenarios as a function of load resistance; and 
         FIG.  51    shows a third embodiment. 
     
    
    
     DETAILED DESCRIPTION 
       FIG.  1    illustrates an energy transfer apparatus  100  coupled to a user&#39;s body  105  according to a preferred embodiment. The energy transfer apparatus  100  comprises a power transmitter  201  in the form of a smartphone coupled to the user&#39;s body  105  and six power receivers  301   a ,  301   b ,  301   c ,  301   d ,  301   e ,  301   f  in the form of wearable or implantable electronic devices also coupled to different parts of the user&#39;s body  105 , via electrodes  300 . Specifically, the power transmitter  201  is coupled to a wrist of the right hand of the user&#39;s body  105 , and a first power receiver  301   a  in the form of earbuds is attached to the head, a second power receiver  301   b  in the form of a smart band-aid is attached the upper left arm, a third power receiver  301   c  in the form of a smart watch is attached to a wrist of the left hand, and a fourth power receiver  301   d  in the form of a glucose sensor and injector is attached to the abdomen, a fifth power receiver  301   e  in the form of an ECG is attached to the chest, and a sixth power receiver  301   f  in the form of a fitness tracker is attached to the left ankle. As illustrated, the six power receivers  301   a ,  301   b ,  301   c ,  301   d ,  301   e ,  301   f  are operable to receive electrical power via an electrical signal transmitted by the power transmitter  201  via capacitive coupling in a power transfer receiving mode of the power receivers  301   a ,  301   b ,  301   c ,  301   d ,  301   e ,  301   f  and to harvest ambient EM waves from ambient electrical sources  101  via electrical signals within the user&#39;s body  105  in a power harvesting mode of the power receivers  301   a ,  301   b ,  301   c ,  301   d ,  301   e ,  301   f , low-frequency ambient EM waves being observed to couple onto the human body, with the 50/60 Hz wave dominating. 
     Thus,  FIG.  1    illustrates a body-coupled power transmission approach with a real-life application scenario, and a body-coupled ambient energy harvesting approach with potential real-life sources. 
       FIG.  2   a    illustrates a simplified representation of the apparatus  100  of  FIG.  1    in the power transfer receiving mode of the apparatus  100 , albeit with only one power receiver  301  and four receiver electrodes  300  shown for clarity. The power transmitter  201  is coupled to the user&#39;s body  105  via a transmitter signal electrode  200 , and the power receiver  301  is coupled to the user&#39;s body  105 , via one of the four receiver signal electrodes  300 . The transmitter  201  and receiver  301  are arranged such that no explicit loop between them exists; the capacitive coupling return path is completed by the parasitic capacitance C RTN    203 ,  303  formed via the ground plane-Earth ground coupling, i.e. the transmitter  201  and receiver  301  both comprise floating ground nodes which are together operable to form the return path for the power transmission via the parasitic capacitances C RTN    203 ,  303  with an external ground. Meanwhile, the parasitic capacitance between the power transmitter ground and the user&#39;s body C LKG    103  and the power receiver ground and the user&#39;s body C LKG    103  results in leakage. Therefore, the apparatus  100  is arranged so that any direct contact between the transmitter  201  and receiver  301  ground planes with the body  105  surface is avoided. 
       FIG.  2   b    illustrates a simplified representation of the apparatus  100  of  FIG.  1    in the power harvesting mode of the apparatus  100 , again with only one power receiver  301  and four receiver electrodes  300  shown for clarity. The power receiver  301  is coupled to the user&#39;s body  105 , via one of the four receiver electrodes  300 . The transmitter  201  plays no role in this mode. 
       FIG.  3    illustrates the architecture of the energy transfer apparatus  100  of  FIGS.  1  and  2    with a single power receiver  301  illustrated as an example. 
     The power transmitter  201  has a power transmitter integrated circuit (IC)  2015  in communication with an off-chip controller  205 . A signal generator in the form of a digitally controlled oscillator (DCO) and driver  207  are integrated onto the IC  2015 . The output impedance of the signal generator  207  is variable by means of a dynamic impedance matching circuit  209  also integrated into the IC  2015  having an electrical signal amplitude detector in the form of feedback module  2029  and two capacitor banks  2011 ,  2013  in electrical connection with an off-chip inductor  2015  to which the electrode  200  is connected, which together determine the power transmitter output interface impedance, as will be explained below. The dynamic impedance matching circuit  209  includes a floating ground node  2051  of the transmitter forming an electrical signal return path to the transmitter  201  via parasitic capacitance  203  with the external ground. The forward transmission path formed by electrode  200  and the return transmission path  203  are therefore connected either side of the inductor  2015 . 
     In this embodiment, the power receiver  301  comprises a power receiver integrated circuit (IC)  305  having a rectifier in the form of a bulk adaptation rectifier  307  and a DC-DC converter in the form of a dual-mode buck-boost converter  309  comprising a switch and pulse generator  3011  and a dual-mode Maximum Power Point Tracking module  3013 . A signal electrode  300  is connected to the power receiver IC  305  via an input impedance circuit in the form of an off-chip detuned impedance booster circuit  3015  comprising a parallel L-C circuit. The impedance booster circuit  3015  includes a floating ground node  3051  of the receiver forming an electrical signal return path from the receiver  301  via parasitic capacitance  303  with the external ground. The forward transmission path formed by electrode  300  and the return transmission path  303  are therefore connected either side of the L-C circuit. 
     In the transmitting mode of the apparatus, an electrical signal is generated by the signal generator in the form of the DCO and Driver  207  and dynamic impedance matching at the environment interface is performed at the TX  201  by the output impedance circuit in the form of dynamic impedance matching circuit  209 . At the RX  301  front-end, the detuned impedance booster (DIB)  3015  and the bulk adaptation rectifier (BAR)  307  are implemented, to perform power recovery enhancement of the signal received via capacitive coupling via electrodes  200 ,  300 . In the case of body-coupled energy harvesting ( FIG.  2   b   ), the DIB  3015  is disconnected, with the BAR  307  still in operation. In both power harvesting ( FIG.  2   b   ) and power transfer receiving mode ( FIG.  2   a   ) modes, the dual-mode buck-boost converter (DM-BBC)  309  performs maximum power point tracking and load regulation at 1.1 V, the nominal supply voltage for standard MOSFETS in 40 nm. 
     The detailed block diagram of the power transmitter  201  is shown in  FIG.  4    with the resistive and capacitive components of environmental impedance resulting from connection with the body  105  represented schematically by resistor capacitor circuit  2023  and includes contributions from both the skin-electrode parasitic capacitance at the signal node  200 , including a reflection coefficient, and the parasitic capacitance at the return path floating node  2051 , both nodes being connected across the inductor  2015 , as shown in  FIG.  3   . The digitally controlled oscillator (DCO)  2071  is arranged to generate a 40 MHz square wave, which outputs to a driver  2072  with an output impedance of 50Ω. A dynamic impedance matching circuit  209  is arranged in between the driver  2072  output and the electrode  200  interface. The dynamic impedance matching circuit  209  has two capacitor banks  2011 , 2013  and an off-chip inductor  2015 . Each capacitor bank  2011 , 2013  has eight capacitors  2027  arranged in parallel, each capacitor  2027  with a corresponding switch  2025  which enables the individual capacitors  2027  to be switched on or off to obtain different configurations of each capacitor bank  2011 ,  2013 . The eight configurations of each capacitor bank  2011 , 2013  are controlled by the controller  205  arranged to output a control signal to the dynamic impedance matching circuit  209 . 
     The first capacitor bank  2011  is arranged in series with the off-chip inductor  2015  and second capacitor bank  2013 . The second capacitor bank  2013  is arranged in parallel with the off-chip inductor  2015 . This arrangement enables the first capacitor bank  2011  to offer an adjustable capacitive element of the dynamic impedance matching circuit  209  and the second capacitor bank  2013  and the off-chip inductor  2015  to offer an adjustable inductive element for L-C impedance matching. 
     The transmitter further includes a ramp generator  2017 , and envelope detector  2021 , with outputs compared by comparator  2019  for input into the controller  205  which includes a digital clock. Together, the ramp generator  2017 , and envelope detector  2021  and comparator  2019  make up the feedback module  2029  shown in  FIG.  3   . 
     In operation, dynamic matching of the environmental impedance (as represented by resistor-capacitor circuit  2023  in  FIG.  4   ) is performed, which automatically configures the capacitor banks  2011 ,  2013 . In conditions of matched impedance (where the driver  2072  output impedance equals the equivalent load impedance represented by  2023 ), half of the open-circuit voltage of the driver  2072  falls across the load. Therefore, an amplitude detector, in the form of an envelope detector (ED)  2021  (see N. M. Pletcher, S. Gambini and J. M. Rabaey, “A 2 GHz 52 μW Wake-Up Receiver with −72 dBm Sensitivity Using Uncertain-IF Architecture,” IEEE International Solid-State Circuits Conference (ISSCC) Dig. Tech. Papers, vol. 51, pp. 524-633, February 2008 for a detailed explanation) is used to track the output amplitude of the driver  2072  (which will be affected by the configuration of the capacitor banks  2011 ,  2013 ). The output amplitude of the driver  2017  is then compared using comparator  2019  with a ramp waveform generated by the ramp generator  2017  for voltage-time conversion using a digital counter in the controller  205 . Such time information represented by the number of counting cycles provides feedback to the digital controller  205  and guides the capacitor bank  2011 ,  2013  reconfiguration direction, until the driver  2072  output amplitude eventually approaches the half-open circuit amplitude. 
     As explained above, the two capacitor banks  2011 ,  2013  have an 8-bit configurability with the least significant bit (LSB) controlling 50 fF resulting in 65,536 (2 8 ×2 8 ) possible capacitor bank  2011 ,  2013  configurations being obtainable. The off-chip inductor  2015  is 2.2 μH. Area  501  of  FIG.  5    illustrates the coverage obtainable by the dynamic matching network  209 , with the corresponding expected environmental impedance range shown by area  503 . 
     A multichotomic search, in the form of a 2-dimensional tetrachotomy-based searching method is employed to determine the final convergence, as illustrated in  FIG.  6   . The potential environment-equivalent resistance and capacitance ranges are represented by a 2-dimensional searching plane, with the vertical axis being the capacitive element and the horizontal axis being resistive. Each point on the plane thus represents one possible environmental impedance. During the first searching cycle  601 , the plane is divided into 4 quadrants  6011 ,  6013 ,  6015  and  6017 , with the capacitor banks  2011 ,  2013  configured to match each centre point  6019  sequentially. Upon deciding the closest match (by comparing the driver  2072  output amplitude against the half-open circuit amplitude and determining which centre point  6019  corresponds to the amplitude closest to this value), its corresponding quadrant is selected for further sub-divisions in further searching cycles  603 , 605 , 607 . This process is repeated until the driver  2072  output amplitude reaches at least a threshold amplitude, in this preferred embodiment namely ±0.15 V of the half-open circuit amplitude, i.e. the magnitude of the difference between the amplitude of the electrical signal and the half open circuit amplitude is less than 0.15V. 
     The maximum number of searching cycles (i.e., plane/quadrant sub-division) is limited to 4, corresponding to 16 times output amplitude evaluation and thus a worst settling time of 16 ms (where the interval in between two evaluations is 1 ms). Assuming slower environmental impedance change (&gt;100 ms, as the human body  105  movement is generally below 10 Hz), this may help to prevent instability concerning the algorithm. 
     A new search is triggered if the feedback circuit  2029  detects that the driver  2072  output amplitude has drifted from the half-open circuit amplitude of 1.25V. A search is triggered whenever the current output amplitude is more than 0.15V away from the half-open circuit amplitude of 1.25V. 
       FIG.  7    shows the measured transient behaviour of the TX  201  output power  707  and the driver  2072  output amplitude  705  during an example searching procedure. In this experiment, a capacitor bank  2011 ,  2013  configuration initially matched under one environmental setting was subject to interface impedance changes, and its driver  2072  output amplitude  705  drifted away from its half-open circuit amplitude (1.25 V) at  701 , thus triggering the searching for a new matched capacitor bank  2011 ,  2013  configuration at  709 . As the search is performed the configurations of the capacitor banks  2011 ,  2013  are altered, as visible in the changing driver  2072  output power  705  and corresponding changes in the TX  201  output power  707 . The amplitude  705  converges to the half-open circuit amplitude of 1.25V at  703  after which no further changes to the capacitor bank  2011 ,  2013  configurations are made. A resulting 7.5 times improvement in the output power  707  is observed upon convergence relative to the power value at the start of the search  709 . 
     Turning now to the power receivers  301 ,  FIG.  8    illustrates the front end of the receiver  301 , including the input impedance circuit in the form of the detuned impedance booster (DIB)  3015 . The impedance booster comprises a parallel L-C circuit  801  comprising inductor  803  and external capacitor  805 . The L-C circuit  801  is arranged so as to be in parallel with the equivalent rectifier input capacitance (represented by  809 ). The resistance of the circuit is defined by load resistance R I  ( 807 ). In this arrangement, the inductor  803  resonates with the overall capacitance of both the external capacitor  805  and the equivalent rectifier input capacitor  809  and the input load impedance of the power receiver  301  is boosted at the resonance frequency of the L-C circuit  801 . The input load impedance of the power receiver  301  includes contributions from both the skin-electrode parasitic capacitance at the signal node  300 , including a reflection coefficient, and at the parasitic capacitance at the return path floating node  3051 , both nodes being connected across the inductor  3015 . The properties of the inductor  803  and external capacitor  805  determine the bandwidth of the resonance response of the L-C circuit  801  which is intentionally detuned, as will be explained below. A switch  811  is provided to enable the impedance booster  3015  to be switched off in the power harvesting mode. 
     According to the preferred embodiment, the inductance of the inductor  803  is 1.2 μH and the capacitance of the capacitor  805  is 6.8 pF, so that the boosted RX  301  input impedance is measured around 8 kΩ, as shown in  FIG.  9   , which shows the receiver  301  input impedance with  903  and without 901 the detuned impedance booster  3015 . The boosted RX  301  input impedance  903  is around the same level as the impedance of C RTN    303  at the resonance/transmission frequency of 40 MHz. 
     The L-C circuit  801  is detuned for impedance matching across a wider range. Detuning is performed to achieve around 6-10×impedance boosting  903  across a 2.8 MHz bandwidth  905 , which has been found to provide the best performance for a transmission frequency of 40 MHz. 
       FIG.  10    illustrates the bulk adaptation rectifier (“BAR”)  307 . The BAR  307  has eight switches  5201 - 5208  which are arranged into two groups  5201 - 5204  and  5205 - 5208 .  5201 - 5204  are arranged as a full-wave gate differential-drive cross-coupled rectifier with applied voltage V IP    5209 -V IM    5211  and output signal V RECT    5213 . Specifically,  5201  and  5202  are p-well transistors gated by V IM    5211  and V IP    5209  respectively with bulk connected towards ground and  5203  and  5204  are deep n-well transistors gated by V IP    5209  and V IM    5211 , respectively with bulk connected towards V RECT    5213 . Switches  5205 - 5208  are n-well transistors arranged to connect the bulk of switches  5201  to  5204  to the applied voltage V IP    5209 -V IM    5211 . Specifically switches  5208 , 5205  are arranged to connect the bulk of switches  5203 ,  5202  respectively to V IP    5209 ; and switches  5206 , 5207  and arranged to connect the bulk of switches  5204 , 5201  respectively towards V IM    5211  The BAR  307  therefore has two modes of operation: switches  5205 - 5208  switched off (“non-bulk biased mode”, or equivalently “normal mode”) and switches  5205 - 5208  switched on (“bulk biasing mode”). 
       FIG.  11    illustrates the operation of the BAR  307  in the two modes (certain reference numerals from  FIG.  10    are omitted in  FIG.  11    for conciseness). When the input amplitude drops below a threshold value, for example 0.4 V, the rectifier  307  is switched to a bulk biasing mode in which switches  5205 - 5208  are turned on as shown by configuration  1003 , which connects the bulk of switches  5201 - 5204  towards V IP    5209  and V IM    5211 , allowing their bulk voltages to alter dynamically based on the AC input. In configuration  1005 , during the half-sine cycle where V IP    5209  is higher than V IM    5211 , switches  5201 ,  5203  are in the forward conduction path and are forward bulk biased. Their threshold voltage is thus reduced, allowing more forward conduction current to V RECT    5213 . On the other hand, switches  5202 ,  5204  are in the leakage path and are reverse bulk biased, with an increased threshold to suppress the leakage current flow out of V RECT    5213 . The same applies to the other sine cycle in configuration  1007 , where V IM    5211 &gt;V IP    5209 . In this mode, the source/drain-bulk DC leakage current due to the bulk potential adjustment is increased, but still  4 - 6  orders of magnitude lower than the conduction current. The more prominent AC leakage due to bulk capacitance (not caused by the bulk adaption proposed) will be directed towards V RECT    5213  before it degrades the power recovery. 
     At higher input amplitude (for example &gt;0.6V), the dynamic Vis is much greater than V TH . Meanwhile, the body diodes start to approach their turn-on voltage and cause increased leakage, making the bulk biasing  1003  no longer necessary nor efficient and the rectifier  307  is switched back to a non-bulk biased mode  1001  in which switches  5205 - 5208  are turned off which reconfigures the bulk of  5201  to  5204  towards V RECT    5213  and ground. This aligns the body diode towards the V RECT    5213 , which effectively avoids leakage via the body diode. Whether the rectifier  307  is in bulk biasing mode  1003  or non-bulk biased mode  1001  is determined by the comparison between V RECT    5213  and a reference voltage, which is 0.4V according to the preferred embodiment, using a duty-cycled dynamic comparator (not shown), i.e. it depends on the amplitude of the electrical signal. 
       FIG.  12    illustrates the operation of the BAR  307  and shows the simulated V SB /V BS  (for PMOS and NMOS, respectively) behaviour using the BAR  307 . V IP    1201  and V IM    1203  are shown in the top graph with an input amplitude of 200 mV along with rectified signal  1205 . At this voltage, the BAR  307  operates in the bulk biasing mode  1003 . The V SB /V BS  is improved for transistors  5201 ,  5202 ,  5203 ,  5204  along the forward conduction path, and decreased for those transistors  5201 ,  5202 ,  5203 ,  5204  along the reverse leakage path as shown for the BAR  307  on results  1205  and  1209  for V SB  and V BS , respectively compared with the corresponding BAR  307  off results  1207  and  1211 . 
       FIG.  13   a    illustrates a simplified architecture of the Dual Mode Buck-Boost Converter (DM-BCC)  309  operating in the discontinuous conduction mode (DCM). The Converter  309  comprises a switch matrix  1301 , a pulse generation module  1303  and a controller in the form of a pulse width controller  1305 , as well as control switch drivers  1307  and an adaptive level shifter (ALS)  1309 . 
     The switch matrix  1301  has two p-well transistors  1311 , 1313  with an inductor  1319  connected in series between them and two n-well transistors  1315 , 1317  connecting either side of the inductor  1319  to ground. The gate voltage of transistor  1311  is controlled by control signals generated by the ALS  1309  with the gate voltage of transistors  1313 - 1317  controlled by control signals generated by the switch drivers  1307 . 
     In operation, the switch matrix  1301  converts input voltage V IN  output by the rectifier  307  and converts it to V OUT  for supplying a device or storage. V OUT  is regulated to a constant 1.1V, which is the nominal supply voltage for standard MOSFETS in 40 nm. 
     The pulse generation module  1303  comprises Tuneable T 1  ( 1325 ), T 2  ( 1323 ) and T 0  ( 1321 ) delay modules, each of which has a plurality of configurations, a fixed delay module  1327  and a NOR gate  1337  with the total delay from all modules  1325 ,  1323 ,  1321  and  1327  and an enabling signal  EN   1335  as inputs, with  EN   1335  held to 0 when half the output voltage V OUT  is below a reference voltage, as will be explained later. 
     In operation, the pulse generation module  1303  is arranged to output a signal PHI 1  to both the ALS  1309  and switch drivers  1307  and the pulse width controller  1305  to output signal PHI 2  to the switch drivers  1307  for control of the switch matrix  1301  and resulting in charging and discharging of the inductor  1319  according to switch  1311 ,  1313 ,  1315 ,  1317  configuration.  FIG.  13   b    shows a schematic representation of the expected PHI 1 , PHI 2  signals and resulting current I L  across the inductor  1319 . PHI 1  includes a pulse  1701  of duration T 1  following a MPPT range settling period, or equivalently discontinuous mode (DCM) time  1703  of the converter  309 , T 0 . PHI 2  includes a pulse  1705  of duration T 2  which occurs after T 1 . T 1  and T 2  correspond to the inductor  1319  charging and discharging time, respectively, as can be seen from the peak  1707  in the current I L . 
     The configurations of the tuneable delay modules  1321  to  1325  are controlled by modules  1325 ,  1329 ,  1331  in the pulse width controller  1305 , specifically Maximum Power Point Tracking (MPPT) module  1329 , which controls the configuration of the tuneable T 1  delay module  1325 ; the Zero Current Switching (ZCS) module  1331  which controls the configuration of the tuneable T 2  delay module  1323 , and the mode selector (MODE SEL)  1333  which controls the configuration of the tuneable T 0  delay module  1321 , and thereby the operating mode of the receiver  201 , as will become apparent from the below. 
       FIG.  14    schematically illustrates the T 1  and T 2  delay modules  1325  and  1323 , respectively of the pulse generation module  1303 . The modules  1325 ,  1323  each comprise a tuneable RC delay line  1401  in parallel with a direct connection  1407  providing inputs to NOR gate  1403  with the input from direct connection  1407  inverted. Tuning of the RC delay line  1401  enables 32 configurations ([0] to [31]) of T 1 /T 2  to be obtained, with exemplary configuration [0]  1405  and configuration [31]  1409  shown in  FIG.  14   . 
       FIG.  15    schematically illustrates the Adaptive level shifter (ALS)  1309  in combination with one of the plurality of switch drivers  1307 . 
     The ALS  1309  comprises a dynamic comparator  1501  which compares half V IN , the input voltage of the converter  309 , against a reference voltage V REF  with V DD  as its supply and PHI 1  as the clock. The dynamic comparator  1501  outputs to a level shifter  1503  and a NOT gate  1509  which are arranged in parallel. The level shifter  1503  has V DD  and V VIRT  as the low and high supplies, and the output  1505  of the level shifter  1503  is applied as a gate voltage to p-well transistor  1507  with supply voltage V DD , The NOT gate  1509  has V DD  as the supply and its output signal is applied as gate voltage to p-well transistor  1511 , with supply voltage V IN . 
     The generation of the gate control signal PHI 1 B_ALS has a NOT gate  1513  taking the PHI 1  signal as input, a level shifter  1515  and the switch driver  1307  giving output PH 1 B_ALS which is the gating signal of switch  1311  of the switch matrix  1301 . Both level shifter  1515  and switch driver  1307  take the maximum V VIRT  of V DD  and V IN  as the supply, V VIRT  being determined by ALS  1309 . Level shifter  1515  also takes V DD  as a lower supply. 
     In operation, the dynamic comparator  1501  triggered by the PHI 1  falling edge compares half V IN  against a reference voltage 1.1 V. V VIRT  takes on the higher potential between V IN  and V DD , and is used as the supply PHI 1  generation. The level-shifter  1503  turns off the header  1507  when V VIRT  is higher than the threshold voltage. Several start-up techniques are known in the art (see, for example, L. Lin, S. Jain and M. Alioto, “A 595 pW 14 pJ/Cycle microcontroller with dual-mode standard cells and self-startup for battery-indifferent distributed sensing,” IEEE International Solid-State Circuits Conference (ISSCC) Dig. Tech. Papers, vol. 61, pp. 44-46, February 2018). 
       FIG.  16    schematically illustrates the tuneable T 0  delay module  1321 . The module  1321  comprises a CMOS inverter  1607  (having n-well transistor and p-well transistors connected in series) connected to V DD  via a circuit comprising a pair of p-well transistors  1601  and  1603  connected in parallel via respective switches  1605  and  1609  and both gated by V B_IREF , which is 0.6V in the preferred embodiment. The output of the CMOS inverter  1607  is connected to static comparator  1619  which compares it with reference voltage V REF . 
     The output of the CMOS inverter is also connected to parallel capacitors  1615 ,  1617  via switches  1611  and  1613 , respectively. 
     In operation, the four switches  1605 ,  1609 ,  1611  and  1613  enable four configurations of the T 0  delay module  1321  (configurations 1 to 4) and thereby four selectable values of T 0   1703 . Selection of T 0   1703  is made according to the operation mode of the converter  309  (i.e. power harvesting or power transmission) and input power in each mode, the converter  309  input power P IN  being lower for harvested energy than for transmitted energy and therefore requiring larger T 0   1703  for efficient conversion. In other words, the mode (i.e. power harvesting or transmission) determines T 0  first (the initial four configurations are narrowed to two configurations). Then depending on the input power in each mode, a final T 0  is chosen. 
     This T 0   1703  is utilized for the converter  309  impedance to settle down fast to the mean Maximum Power Point (MPP), after which the impedance monitor loop (using T 2  as the feedback) adjusts T 1  dynamically to fine-tune the MPP, with the step size of 12 ns and 32 steps in total in this preferred embodiment. 
     The effect of T 0   1703  and T 1  on the converter  309  impedance is expressed by: 
         Z   DM_BBC =2 L·T   SW   /T 1≈2 L·T 0/ T 1  (1)
 
     where L is the inductance and T SW  is the switching period. T 0   1703  dominates the T SW  by being larger than T 1  and T 2  by ˜7-4000λ in this preferred embodiment, and thus the approximation above. The relationship between P IN  and T 2  for the buck-boost converter  309  is given by 
     
       
         
           
             
               
                 
                   
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     Therefore, T 2  is positively proportional to P IN . Employing the assumption that V IN  varies slower than 20 ms (10 T SW_MAX ), with V OUT  and L fixed, and T 0   1703  maintained the same for each configuration, T 2  then serves as the indicator of the P IN  level to guide the T 1  adjustment direction for MPP fine-tuning. 
     The converter  309  operates asynchronously to suppress quiescent current. The pulse generation involving T 0   1703 , T 1 , and T 2  self oscillates, as long as V OUT  drops below the threshold voltage, for example 1.1 V (with  EN   1335  held to 0, which is decided by a dynamic comparator triggered by the PHI 1  falling edge) which indicates the need of power delivery to the load. The maximum conversion frequency f SW  is 1/(T 0 +T 1 +T 2 ), where T 0   1703  dominates. The conversion frequency is adaptively reduced under lighter loading conditions, with the pulse generation halted upon V OUT  charged over the threshold voltage 1.1 V. T 1  and T 2  are generated by tuneable RC delay lines  1401 . While the duration of T 1  is determined by the MPPT module  1329 , the T 2  duration is decided by the zero current switching (ZCS) module  1331  (see L. Lin, S. Jain and M. Alioto, “Integrated Power Management for Battery-Indifferent Systems With Ultra-Wide Adaptation Down to nW,” IEEE Journal of Solid-State Circuits, vol. 55, no. 4, pp. 967-976, April 2020 and D. EI-Damak and A. P. Chandrakasan, “A 10 nW-1 μW Power Management IC With Integrated Battery Management and Self-Startup for Energy Harvesting Applications,” IEEE Journal of Solid-State Circuits, vol. 51, no. 4, pp. 943-954, April 2016 for a description of ZCS). T 0   1703  is generated by the tuneable charging of capacitors  1615  and  1617  by a leakage-based current reference. The Adaptive Level Shifter (ALS)  1309  is employed so that PM 1  can be fully turned off when V IN  is greater than the threshold voltage 1.1 V. 
     The dual-mode buck booster converter  309  therefore operates in the following way. First, T 0   1703  is selected according to an operating mode of the converter  309  (corresponding to the mode of operation of the receiver  301 ) and corresponding input voltage and power; i.e. the most suitable configuration of the T 0  module  1321  given the energy source (i.e. ambient or transmitted energy) and voltage is controlled by mode selector  1333 . Two of the T 0   1703  configurations are employed in a power harvesting operation mode of the converter  309  and two of the T 0   1703  configurations are employed in a power transmission operation mode of the converter  309 , i.e. there are two predetermined values of T 0   1703  for each mode, with each of the two predetermined values corresponding to optimal converter  309  efficiency in different input voltage ranges. 
     The mode selector  1333  checks the accumulated voltage on a storage capacitor at the converter  309  input, determining periodically if there is transmitted power (indicated by the input voltage). If transmitted power is available, it remains in the transmission mode, otherwise it scavenges power by switching to the harvesting mode. The mode selector  1333  further compares the input voltage of the converter  309  with a pre-defined voltage reference in order to determine which value of T 0   1703  to employ for the given mode. 
       FIGS.  17   a  and  17   b    illustrate the efficiency of the converter  309  according to the preferred embodiment as a function of switching frequency f SW , with  FIG.  17   a    showing power transmission and  FIG.  17   b    showing power harvesting. In  FIG.  17   a   ,  1701  indicates 1V input voltage,  1703  indicates 0.7V,  1705  indicates 0.5V and  1707  indicates 0.3V. In  FIG.  17   b   ,  1709  indicates 1.5V,  1711  indicates 0.8V and  1713  indicates 0.4V.  FIGS.  17   a  and  17   b    show that the optimal switching frequency f SW  increases under higher input voltage, and vice versa. This is because at the rectifier  307  load (converter  309  input impedance Z DM_BBC ), the optimal impedance decreases at higher input voltage (due to higher I ON  and lower rectifier  307  impedance). This then leads to higher current into the converter  309  and thus higher conduction loss. 
     Based on the measurements shown in  FIGS.  17   a  and  17   b   , the T 0  1703 value corresponding to the switching f SW  (assuming f SW =1/T 0 ) closest to the optimal switching frequency for the particular operation mode of the converter  309  (transmission or harvesting) and input voltage to be received is selected. In practice, this means that T 0   1703  configurations 1 and 2 are therefore selected in the power transmission operation mode, with configuration 1 being employed at input voltages greater than 0.55V and configuration 2 being employed at voltages less than or equal to 0.55V. Configuration 1 corresponds to switches  1605 ,  1609  and  1611  being set to closed and  1613  to open, while configuration 2 corresponds to all four switches  1605 ,  1609 ,  1611 ,  1613  closed. These two configurations give rise to switching frequencies of 280 kHz (configuration 1) and 15 kHz (configuration 2), respectively, calculated assuming f SW ≈1/T 0 . 
     In contrast, T 0  configurations 3 and 4 are selected in the energy harvesting operation mode, with configuration 3 being employed at input voltages greater than 0.55V and configuration 4 being employed at voltages less than or equal to 0.55V. Configuration 3 corresponds to switches  1605  &amp;  1611  being set to closed with the other switches open, while configuration 4 switches  1605 ,  1611  &amp;  1613  closed. These two configurations give rise to switching frequencies of 6 kHz (configuration 3) and 500 Hz (configuration 4), respectively, calculated assuming f SW ≈1/T 0 . 
     Following T 0   1703  selection, the inductor  1319  charging time T 1  which corresponds to the maximum input voltage to the DC-DC converter  309  is determined according to equation (1) within the range 100-480 ns, enabling the target Z DM_BBC  span to be covered. The switch sizing is also optimized by balancing the switching and conduction loss, with larger switches lead to higher switching loss but lower conduction loss, and vice versa. 
       FIG.  17   c    shows an outline of the method of determining T 1  and T 2 . In step S 5301 , the T 0   1703  configuration is selected using the mode selector  1333 . In step S 5303 , an initial T 1  configuration is selected. In S 5305 , the ZCS configuration of T 2  is determined given the selected T 0   1703  configuration and the T 1  configuration. In S 5307 , the T 1  configuration is altered to an adjacent configuration. The direction of adjustment of the T 1  configuration is determined by the trend of change in the value of the inductor discharging time (T 2 ) which represents the input power trend of change, as will be described below. The method then returns to step S 5305 . 
       FIG.  17   d    illustrates the method of asynchronous MPPT performed to determine T 1 , i.e. the method performed in step S 5307  of  FIG.  17   c   . Upon T 2  being updated by ZCS in step S 5305 , the T 2  register is updated in  51801  and the completion status of T 2  adjustment under a given T 1  is determined in  51803 . If the T 2  is settled, in step S 1805  it is determined if the converter  309  is operating at the maximum frequency f SW . If both of these requirements are met, in step S 1807  the T 1  register is updated, i.e. it is shifted by one bit in the direction determined in S 5307 , and in step S 1809 , the currently settled T 2  is compared with the stored value and the T 1  increment/decrement direction is determined and stored. If the T 2  settled value is larger than the earlier settled value, T 1  is incrementally adjusted in the same direction as in the previous adjustment. Otherwise, T 1  adjusted in the opposite direction. At the very start of the process, T 1  is incrementally increased, introducing perturbation for T 2  change, which then affects the T 1  direction later on. In step S 1811  the current T 2  value is stored, and in step S 1813 , the ZCS decision is stored, i.e. the incremental change in T 2 . After the method of  FIG.  18    has been performed, a cycle check is performed and ‘T 2  unsettled’ is assumed at the following cycle, should this be the first T 2  value under the newly changed T 1 . At the steady state, the T 1  configuration will oscillate in between two adjacent values which are upper and lower boundaries of the optimal value (yielding maximum power). 
     As explained above, the apparatus  100  according to the preferred embodiment has two modes of operation: 1) a power transfer receiving mode, as shown schematically in  FIGS.  2   a   , and 2) a power harvesting mode, as shown schematically in  FIG.  2     b.    
     By default, the apparatus  100  operates in the power transmission operation mode. In this mode, the transmitter  201  is switched on and power is generated at the DCO and driver  207 , with dynamic impedance matching, equivalently impedance adjustment, performed by the dynamic impedance matcher  209  prior to transmission via the body  105  via electrode  200 . In the receiver  301 , switch  811  is set to the closed position (see  FIG.  8   ), so that the DIB  3015  is connected and impedance boosting, equivalently impedance adjustment is performed at the receiver  301  front-end in order to help maximise the power of the electromagnetic signal received from the transmitter  201  via electrode  300 . The converter  309  operates in its power transmission operation mode and the mode selector  1333  sends a control signal to the T 0  delay module  1321  to switch its configuration to configuration 1 or 2. The receiver  301  recovers the DC power of the impedance boosted signal via rectification using the BAR  307  and voltage conversion using Dual-Mode Buck-Boost Converter  309  as described above and supplies it to wearable applications or accumulates it at a storage capacitor. 
     The accumulated voltage on a storage capacitor at the converter  309  input is monitored periodically and, if no transmitted power (as indicated by the voltage) is detected, then mode switching is determined and the apparatus  100  switches to power harvesting mode, i.e. mode switching occurs in dependence on a power level of the received electrical signal. 
     In this mode, in the receiver  301 , switch  811  is set to the open position (see  FIG.  8   ), so that the DIB  3015  is disconnected as impedance boosting is not required in this mode. The converter  309  operates in its power harvesting operation mode and the T 0  delay module  1321  is set to configuration 3 or 4 as controlled by mode selector  1333 , the only electromagnetic (EM) field present in the body is due to the coupling of the ambient electromagnetic waves present in the environment due to ambient sources  101 . The receiver  301  picks up the EM field via electrodes  300  at its input and recovers the DC power via the BAR  307  and Dual-Mode Buck-Boost Converter  309  as described above and supply it to wearable applications in which the receiver  301  is integrated or in connection with or, alternatively, the DC power is accumulated. 
     It follows that apparatus  100  according to the preferred embodiment may utilize human body-coupling to deliver power efficiently to the entire body  105 , while at the same time be capable of efficient harvesting energy from ambient EM waves without the need of a bulky antenna or being susceptible to the body shadowing effect. As experimental results discussed below show, the performance of apparatus  100  according to the preferred embodiment may ensure energy transfer independent of the network scale (i.e. the number of receivers  301 ) and enable flexible electrode  300  sizing and electrode  300  location without compromising on energy transfer efficiency. 
     Thus, the preferred embodiment may efficiently address the issue of the powering up of wearables. Using the preferred embodiment, one may be able to power up all the wearables around the body  105  through 1) a single battery source (such as smart phone) transmitting energy through the body  105  (body-coupled active energy transfer), or 2) ambient energy harvested through the body  105  (passive energy harvesting). The ambient energy generated from many ambient sources  101  such as electric devices and their power lines may be coupled with the human body  105 . This coupled energy may deliver to multiple power receivers  301 , and each power receiver  301  may save the energy. In the case of insufficient ambient energy, an active power transmitter  201  may transmit body-coupled energy to the power receiver  301 . 
     The apparatus  100 , may enable powering of multiple wearable or implantable devices by charging only a single device, such as a smartphone, and delivering power to the wearables using that single device; individual charging of all of the wearables is not required. Further, by switching to harvesting mode the battery lifetime of the devices may be extended even when the batteries of the power transmitting device have been depleted. 
     Apparatus  100  according to the preferred embodiment may therefore enable the battery lifetime or help to sustain the fully autonomous operation of wearable sensing technology or other devices. A number of features of the preferred embodiment may contribute to the usefulness of apparatus  100  according to this embodiment. These will now be discussed in detail. 
     Firstly, dynamic impedance matching by the dynamic impedance matcher  209  at the TX  201  output using dynamic impedance boosting may enable the apparatus  100  to maintain optimal power transmission amidst varying environmental impedance variations at the TX  201  output interface and therefore help to ensure robustness of the power transmission in light of environmental changes. TX  201  output interface consists of two nodes—ground node  2051  (forming the return path with a parasitic capacitance against the Earth ground), and the signal output node (electrode  201 ) forming the forward conduction path with skin-electrode parasitic capacitance and body channel capacitive coupling. The impedance matching according to the preferred embodiment is introduced at the signal node with ground node  2051  floating. Therefore, it may account for the overall effect of all parasitic coupling at both the ground node  2051  and the signal node (electrode  201 ), and may therefore improve the reflection coefficient and thus power transfer efficiency. 
     As discussed above, the TX  201  output interface is unconventional, with the load impedance contributed by the skin-electrode  300  contact and the human body  301  path impedance in its forward transmission direction, as well as the ground plane-Earth ground parasitic capacitance  203 , 303  in its return path. Thus, the S 11  parameter of the transmitter  201  will vary due to the impedance variations induced by posture, environment, individual and electrode  300  contact changes. This is illustrated in  FIG.  18    which illustrates the impedance variation according to environmental conditions. Line  1901  is the S 11  parameter of −23 dB achieved by L-C impedance matching at the transmission frequency of 40 MHz, on subject in a sitting position wearing a wet electrode. In contrast, line  1903  shows S 11  for the same subject in a squatting position and line  1905  in a standing position, lines  1907 , 1909  show S 11  for two different environmental scenarios, lines  1911 , 1913  for two different subjects, and lines  1915 , 1917  for half dry and fully dry electrodes, respectively (without rematching of the impedance). As can be seen from  FIG.  18   , due to the impedance variations induced by posture, environment, individual and electrode  300  contact changes, the S11 behaviour is changed and is observed to degrade by up to 20 dB at 40 MHz. Considering the TX  201 -body  105 /environment as a two-port network, with the output interface (incorporating both the forward transmission path and the return path impedance) measured as a whole, the resistive component is expected to range from 2 kΩ to 30 kΩ, and the capacitive component from 0.1 pF to 7 pF, as shown by area  503  in  FIG.  5   . 
     By employing dynamic impedance matching according to the preferred embodiment, apparatus  100  according to the preferred embodiment may enable S 11  and therefore power to be enhanced for all environmental conditions. As can be seen in  FIG.  5   , the impedance range achievable by apparatus  100  according to the preferred embodiment, indicated by area  501 , encompasses all of the expected impedance variation due to environmental conditions, indicated by area  503 . 
     Measurements of power recovery using apparatus  100  according to the preferred embodiment were performed on the human body, with wet electrodes  200 ,  300  (Red dot, 3M Ag/AgCl,  2237 ) used as the interface to couple the electric field onto and from the skin surface. The TX IC  2015  and the RX IC  305  were fabricated using 40 nm CMOS technology, with the TX IC  3015  occupying 0.21 mm 2  and the RX IC 305 0.29 mm 2 . Off-chip components required for the design include the inductor  3015  used for TX  201  impedance matching, the L-C tank (1.2 μH, 6.8 pF) for DIB  3015  at the RX  301  front-end, and the converter inductor  1319  (47 μH). The TX  201  was battery powered, and a USB-powered picoscope was employed to avoid additional return path coupling induced by equipment. Both the TX  201  and RX  301  PCB were 4×4×1 cm 3 . 
       FIG.  19    shows the recovered powered with dynamic impedance matching at the transmitter  201  (shown by  2001 ) and without dynamic impedance matching at the transmitter  201  (shown by  2003 ) over a range of distances. The inset shows an expanded view for the measurements at larger distances. As can be seen from  FIG.  19    at distances over 30 cm, no power is recovered without dynamic impedance matching at the transmitter  201  and dynamic impedance matching at the transmitter  201  enables an improvement in power recovery of more than 3 times at smaller distances. Overall, the power coverage achieved four times the range with the dynamic impedance matching at the transmitter  201  than without any impedance matching. 
     Additionally, the capacitor bank  2011 ,  2013  design of the dynamic impedance matcher  209  avoids the usage of an inductor bank, thereby facilitating on-chip integration. 
     By employing the Bulk Adaptation Rectifier (BAR)  307  which may enable bulk biasing of the rectifier  307  at low voltages in the receiver of the proposed embodiment, power recovery at low voltage may be improved, which in turn may further enhance power recovery over transmission at longer distances by increasing the sensitivity of the device.  FIG.  20    shows results for the recovered power with input amplitude using a receiver  301  according to the preferred embodiment with the BAR  307  turned off  2005 , turned on  2007  and the ratio  2009  between the two results. The ratio  2009  shows that an almost 4 times improvement was obtained at an input amplitude of 0.2V. This may enable power to be transmitted at longer distances over the body  105 . 
     The high sensitivity potentially conferred by the BAR  307  may also enable harvesting of the low frequency ambient EM waves (&lt;hundreds of kHz, for example the 50/60 Hz EM wave from power lines) coupled to the body which may not be possible to harvest using antenna-based approaches due to impractical antenna sizing. 
     The detuned impedance booster  3015  may also contribute to improved performance at longer distances by increasing the voltage received at the receiver  301  and therefore improving the rectifier  307  performance. The parasitic capacitance C RTN    303  formed by the ground plane to the Earth ground coupling completes the circuit loop and contributes positively to power recovery. Generally, below 1 pF, the C RTN    303  is the largest environmental impedance drawing away voltage and power from the RX  301  input, whereas others such as skin-electrode interface exhibits ˜100 nF parallel capacitance (using a wet electrode 3M  2237 ) and an overall impedance of ˜130Ω at 40 MHz. As shown in  FIG.  9   , the RX  301  input impedance  901  was measured to be ˜600Ω at 40 MHz (˜10 times lower than the fF-level C RTN    303 ), which may result in low input voltage and power observed at the RX  301  front-end. On top of possible degraded power transmission efficiency, the low input voltage may also lead to low rectification efficiency due to the threshold drop. The detuned parallel L-C booster (parallel) circuit is thus introduced for capacitive power transfer circuit, with one node floating. The RX input impedance may therefore be boosted such that higher input voltage/power may be seen at the RX  301 , due to the voltage division against a small return path parasitic capacitance (which translates to large impedance). 
     Boosting the input impedance according to the preferred embodiment may therefore not only increase the voltage received at the RX  301 , but also may enhance the impedance matching for more efficient power transmission. In addition, detuning of the L-C tank of the Impedance Booster  3015  as described above may enable impedance matching across a wider range. This may improve the robustness of the receiver  301  against capacitance variations due to environmental, individual, or setup changes. 
     Thus, the use of impedance matching and boosting may confer three main benefits. First, with the impedance boosted, higher voltage may be observed at the receiver  301  input due to voltage division, which enhances the power available to the receiver  301 . Second, as the voltage across the rectifier  307  increases, the rectification efficiency may increase correspondingly, leading to an overall boost in the end-to-end efficiency Third, the detuned matching may allow for wider impedance tolerance, mitigating the influence of the varying interface/environmental impedance. 
       FIGS.  21   a - 21   c    show the performance of apparatus  100  according to the preferred embodiment with transmission distance to illustrate the effect of both the BAR  307  and DIB  309 .  FIG.  21   a    shows an arm-arm path,  FIG.  21   b    shows results for an ankle-forehead path and  FIG.  21   c    shows results for a path around the waist.  2101  indicates the recovered power using a receiver  301  and transmitter  201  according to the preferred embodiment (i.e. with both BAR  307  and DIB  3015 ).  2103  indicates the recovered power using a transmitter  301  according to the preferred embodiment with a receiver having the DIB  3015  but a conventional differential drive rectifier.  2105  indicates the recovered power using a transmitter  201  according to the preferred embodiment with a receiver having only a conventional differential drive rectifier. 
     In these results, both the BAR  307  and DIB  3015  were seen to extend the range of power transmission for all of the paths shown. Specifically, with the TX  201  placed on the subject&#39;s wrist outputting ˜3 mW, the RX  301  power recovery at 15 cm apart along the arm was improved from 1.3 μW to 100 μW using apparatus  100  according to the preferred embodiment, as compared to the power transmission using the TX  201  and a conventional differential drive rectifier of the same size, as can be seen in  FIG.  21   a   . Meanwhile, at ˜120 cm apart where the RX  201  node was placed on the other wrist, 1 μW could still be recovered, extending the coverage by 8 times. The power recovered at 30 cm was improved from 650 nW to 45.2 μW, and the coverage from 30 cm to 160 cm. The end-to-end efficiency from the TX  201  output to the RX  301  recovered power was calculated to be 3.3% at 15 cm apart and 0.03% at 160 cm apart. Along the non-line-of-sight path (non-LoS) of around the waist ( FIG.  21   c   ), the body-coupled approach delivered 5 μW from the front to the back of the body  105 . 
     The variable T 0   1703  (discontinuous conduction mode time) of the converter  309  may enable efficient power recovery in both power transmission receiving and power harvesting modes of the preferred embodiment without an excessive inductor  1319  charging time as will now be explained. 
       FIG.  22    illustrates the optimal RX  301  loading variations under different power transmission conditions (e.g., different on-body transmission distances) or environmental variations, measured with the rectification front-end according to the preferred embodiment, specifically transmission over 30 cm  2201 , transmission over 90 cm  2203 , transmission over 120 cm  2205  and harvesting  2207 . For the body-coupled power transmission ( 2201 ,  2203  and  2205 ), the optimal load resistance was measured to range from 10 kΩ to 50 kΩ, depending on distance. For the body-coupled ambient energy harvesting ( 2207 ), the optimal resistance was observed to fall in a distinct range of around 150 kΩ to 300 kΩ. 
     By employing different discontinuous mode time (DCM) T 0   1703  values according to how the apparatus  100  is being employed therefore, it may enable the impedance loading of the receiver  301  to accommodate and be fined-tuned for optimal power transmission over the full range of 10 kΩ to 300 kΩ required for efficient power recovery in both power transmission configurations and power harvesting configurations without an excessive increase in T 1  (the inductor  1319  charging time) which may degrade the resolution and the settling time during the mode switching. 
     Thus, apparatus  100  according to the preferred embodiment may further enable efficient harvesting of ambient energy by enabling the harvesting of energy via body coupling in addition to power receiving as illustrated above. As shown in  FIG.  23   , the low-frequency ambient EM wave ( 2503 ) is observed to couple onto the human body ( 2501 ). Indeed, up to 30 dB higher EM strength may be achieved with the body-coupling ( 2501 ), as compared to that measured in the air ( 2503 ), with the 50/60 Hz wave dominating. Using the body-coupling mechanism to harvest these body-coupled ambient EM waves in different daily environments, up to 2.5 μW was scavenged experimentally using the apparatus  100  of the preferred embodiment, as illustrated in  FIG.  24    for a range of environments.  FIG.  25    illustrates that the amount of power harvested is indifferent to where the RX  301  electrode  300  was placed on the human body  105  despite slight fluctuations. Thus, apparatus  100  according to the preferred embodiment may enable placement-independent energy scavenging. Thus, simultaneous energy scavenging may be performed by all wearable nodes, thereby increasing total power recovery and the number of wearable devices that may be charged simultaneously. Further the energy harvesting is performed without the use of a bulky antenna. Moreover, unlike conventional harvesters with specific placement requirements to meet, energy harvesting using apparatus  100  according to the preferred embodiment is applicable across the human body  105  regardless of clothing coverage or body blockage. 
     The power consumption of the converter  309  is shown in  FIG.  26    with the breakdown of the converter  309  power consumption illustrated in  FIG.  27   . As T 0   1703  increases at lower input power, the total power consumption is measured to be 7.14 μW, 486 nW, 234 nW, and 94 nW under the 4 T 0   1703  configurations of the preferred embodiment, respectively. The reduction is contributed by both decreased switching activities, and the lower bias current for continuous-time comparators due to higher delay tolerance at larger T 0   1703 . The percentage of pulse width controller (PWC) power takes the dominance as its active power gradually approaches its quiescent power of ˜60 nW, whereas the driver  2072  and pulse generation power still see a decrease (with a quiescent power of 17 nW). As compared to the recovered power range of &gt;30 μW and 800 nW-30 μW for power transmission, and &gt;1 μW and 130 nW-1 μW for the harvesting, the converter  309  may therefore also enable conversion with affordable power consumption. 
     Thus, apparatus  100  according to the preferred embodiment may offer power sustainability at all body locations and enable an interconnected body area powering platform which could support reliable and self-sustained wireless body area electronics. 
     The preferred embodiment should not be construed as limitative. For example, the signal generator (DCO)  2071  may generate a differently shaped wave or a wave of different frequency, instead of a 40 MHz square wave, in particular a wave in the range 30 MHz to 80 MHz, which is outside of the quasi-static field and may exhibit the low path loss and thereby enable delivery and recovery of power across the entire human body. The driver  2072  may have an output impedance other than 50Ω. The dual mode buck-boost converter  309  may regulate the output voltage to a voltage other than 1.1V. The inductance of the inductor  803  may be different than 1.2 μH and the capacitance of the capacitor  805  may be different than 6.8 pF. The inductance of the inductor  803  and the capacitance of the capacitor  805  may be such that the RX  301  input impedance is boosted at a value other than 8 kΩ. The reference voltage for determining the operating mode of the BAR  307  may be higher or lower than 0.4V. 
     While it is preferred for the BAR  307  to have two modes of operation (i.e. bulk biasing mode and non-bulk biasing mode) and operable to selectively switch between the two modes depending on the amplitude of the rectified signal, it is envisaged that it is possible that the BAR  307  may be adapted to operate only in the bulk biasing mode during reception. 
     The bandwidth  905  of the impedance boosting may be greater or less than 2.8 MHz. In particular, the bandwidth  905  of the impedance boosting may be 20% greater or smaller than the L-C central resonant frequency. The bandwidth may be varied in accordance with the transmission frequency or the general environment return path. For example, if the general environment return path impedance is too high, the L-C peak impedance may be boosted to receive more power from environment, at the cost of narrowing down the L-C bandwidth, i.e. decreasing the “detuning” of the input impedance L-C circuit. 
     The capacitive component of the input impedance L-C circuit may comprise one or more actual capacitors, such as capacitor  805  of the receiver  301  of the preferred embodiment, or the capacitance may be generated using other means, for example by the parasitic input capacitance of the subsequent module in the receiver, for example the rectifier, i.e. without the explicit use of a capacitor as a component of the input impedance L-C circuit. Alternatively, the capacitive component of the input impedance L-C circuit may be defined by a combination of the two, i.e. the capacitive component of the input impedance L-C circuit may have contributions from both a capacitor such as capacitor  805 , as a component of the circuit, and, for example, a parasitic input capacitance of another circuit or component in the receiver, such as rectifier input capacitance  809 , as in the receiver  301  according to the preferred embodiment. 
     Instead of switching from the power transmission operation mode to the harvesting mode on the basis of transmitted power received, the apparatus may instead, or in addition, be configured to switch from the harvesting mode to the power transmission operation mode according to the power of the electrical signal received in the harvesting mode. 
     The T 0 , T 1  and T 2  delay modules ( 1321 ,  1323  and  1325 , respectively) may have a fewer or greater number of configurations. For example, the T 0  delay module  1321  may have only two configurations: one for the power harvesting mode and one for the power transfer receiving mode, and the mode selector  1333  may only allow selection between these two modes. 
     In the receiver  301 , the dual-mode buck boost converter  309  may be employed with a rectifier which is not a bulk adaptation rectifier  307 , for example a full-bridge rectifier with a store capacitor. Alternatively, the bulk adaptation rectifier  307  could be employed with a DC-DC converter which is not a dual-mode buck-boost converter  309 , for example a buck-boost converter or a low drop-out regulator. The detuned impedance booster  3015  may be omitted entirely, or the level of tuning (i.e. the bandwidth) of the impedance booster  3015  may be varied. The receiver  301  may be configured to only operate in either the power receiving mode or the power harvesting mode and the load impedance, or load impedance configurations of the converter  309  may only be configured to accommodate one of these modes. Different receivers  301  may be configured to perform energy harvesting and power receiving. 
     A new search for the capacitor bank  2011 ,  2013  configurations may be triggered when the amplitude of the electrical signal is closer to or further from the half open circuit amplitude than 0.15V. The threshold amplitude for the search may be greater or less than ±0.15 V of the half-open circuit amplitude. The threshold amplitude may not be defined with respect to the half-open circuit amplitude, instead being a predetermined minimum amplitude value for the electrical signal. 
     The transmitter  201  according to the preferred embodiment may be employed with a receiver of a different design. The receiver  301  according to the preferred embodiment may be employed with a transmitter of a different design, for example as shown in  FIG.  28   , in which a transmitter  2801  without dynamic impedance matching is employed. One or more of the transmitter  201  and the receiver  301  may not comprise an IC chip  2015 ,  305 . 
     Some modules which are shown in the preferred embodiment as being components of other modules on the IC chip  2015 ,  305 , may in fact be separate modules or realised off-chip. For example, the ALS module  1309  may form part of the switch and pulse generation module  3011  or it may be a separate module, as shown in  FIG.  28   . 
     The dual-mode buck boost converter  309  may not perform T 1  fine tuning, the only impedance load variation in the converter being achieved by varying T 0   1703 . 
     The receiver  301  may comprise a data transmitter and the transmitter  201  may comprise a data receiver, for use with body area network (BAN) for data transmission between them. Different frequencies may be employed for power and data communication to prevent confusion. In this way the transmitter  201  may use the power receiver  301  information and state, such as a rectifier  307  output voltage, to transmit at improved power efficiency. 
     The receiver  201  may include a control module for controlling the modes of the dual-mode buck boost converter  309  (i.e. the T 0  configuration) via the mode selector  1333  and/or the operating mode of the BAR  307 . The receiver controller module may sense the output voltage of the BAR  307  and may send a signal to the power TX  201  to achieve maximum power efficiency, in the case that the receiver  301  may include a data transmitter. It may also generate the control signal for the dual-mode buck boost converter  309  and change the matching network  3015  configuration depending on the situation. 
     The matching network comprised within the DIB  3015  may be modifiable depending on the human subject, for example, by varying the capacitance of its capacitor. 
     The DCO  2071  may employ a high threshold MOSFET for leakage current reduction. 
     The transmitter controller  205  may control all sub-blocks of the Power TX  201 . Based on the data from the RX  301  via BAN, the controller  205  may determine all the parameters and generate a control signal such as frequency and a maximum current of power amplification for safety. 
     The transmitter  201  may comprise a current limiter, which may set the maximum current consumed in the driver  2072 , limiting the maximum power transfer. A binary-weighted current mirror may be used to control the current level digitally. This may help to ensure safety with the transmitter  201  is being employed. 
     The receiver  301  may comprise a storage capacitor for storing the received power, for example in the case that it is not being used immediately for powering a wearable device. 
     The multichotomic search performed by the dynamic impedance matcher  209  may divide the 2-D searching area by greater or fewer sections, for example, the search may be a dichotomy-based search. 
     Although the described preferred embodiment is directed to power transfer using capacitive coupling via the body, power transfer may alternatively be achieved by alternative mechanisms employing the body as a coupling medium. For example, magnetic resonance coupling between power transmitter  201  and receiver  301  may be achieved by employing a pair of coupled inductor coils as electrodes  200 ,  300 . 
     Power transfer could alternatively be achieved via a galvanic current. In this approach, instead of floating ground nodes  2051 ,  3051 , the return path for both transmitter  201  and receiver  301  would be formed by further, respective return electrodes coupled to the body, i.e. two electrodes would be employed at the transmitter  201  output interface and two electrodes and the receiver  301  input interface to form the form the forward and return paths, both via the body  105 , galvanic current between the two sets of electrodes injecting AC current into the skin or body tissue. 
     Although the description of the apparatus of the preferred embodiment has been directed primarily to the use of the apparatus to power wearables, which may involve power transmission primarily (although not exclusively) via the human body surface, i.e. the skin surface, the apparatus may be instead be employed to power implantable devices via capacitive coupling, magnetic resonance coupling or galvanic current, in which case the power transmission may be primarily (although not exclusively) through the body via subcutaneous or other tissue. 
     Further Embodiments 
       FIG.  29    illustrates schematically an energy transfer apparatus  2900  according to a second embodiment for powering wearable devices via energy transfer via the body and energy harvesting. The device comprises a transmitter  2901  and receiver  2903  connected to the body  105  via electrode  6200  and  6300 , respectively, as before in the preferred embodiment. 
     The transmitter  2901  comprises a power source  2911 , a regulator  2909 , an oscillator  2907  and an L-structured matching circuitry  2905  comprising a capacitor  2913  and  2915  to match the impedance between the oscillator  2907  output and the transmitter  2901 -body  105 /environment interface. 
     The receiver  2903  comprises an L-C impedance booster  2917  shunt across the receiver front end, a diode rectifier  2919 , a storage capacitor  2902 , and a resistive load (e.g. that generated by the application of the wearable device)  2921 . 
     The value of the resistive load  2921  is different according to whether the receiver  2903  is to be employed as a power transfer receiver or a power harvester. The L-C impedance booster  2917  is disconnected from the circuitry via switch  2920  (or may be omitted entirely) when the receiver  2903  is to be employed for energy harvesting. 
     In operation, an EM field in the megahertz range is generated by the oscillator  2907  powered by the regulated supply voltage  2911  and coupled onto the human body  105  surface via electrode  6200  which is wet in the described embodiment. The ground node of the transmitter  2901  is floating, which forms a parasitic capacitance C RTN1    2923  with the Earth ground for the return path completion. Meanwhile, the parasitic capacitance between the transmitter  2901  ground and the human body C LKG1    2925  results in leakage. Therefore, as with the preferred embodiment, any direct contact between the transmitter  2901  ground plane with the body  105  surface is avoided. The receiver  2903  picks up the EM field via electrode  6300  at the input and recovers the DC power via a rectification circuitry  2919 . The DC power is then accumulated at the storage capacitor C STR    2902  to offer power supply to wearable applications  2921 . The return path is closed by the parasitic capacitance  2927  in between the receiver  2903  ground node and the Earth ground. Any capacitance (e.g., C LKG2    2929 , C LKG3    2931 ) coupling circuit nodes to the human body  105  contributes to power leakage. 
     To improve the amount of the power recovered, impedance matching is performed at the transmitter  2901 -body  105  interface, via the L-structured matching circuitry  2917  to match the impedance between the oscillator  2907  output and the transmitter  2901 -body  105 /environment interface. 
     Apparatus  2900  according to the second embodiment may enable power receiving through the body  105  to be achieved at wearable devices using only commercially available components, thus presenting immediate potentials in reproduction and integration in wearable devices. 
     Apparatus  2900  according to the second embodiment, shown in  FIG.  29    provided the experimental set up necessary to characterise the usefulness of the preferred embodiment. 
     For characterising the power transfer mode, a transmitter  2901  according to the second embodiment ( FIG.  29   ) was fabricated on CB (0.4 mm thickness, Au electroplating pad) consisting of a battery pack (3/6 V), a regulator (Microchip, MCP1700T-3302E/TT), crystal oscillators (ALE), and matching circuitry (SMD type) at its output. It coupled power via a wet electrode  6200  (Red dot, 3M Ag/AgCl,  2237 ) onto the human body  105 . A receiver  2903  according to the second embodiment received the signal via the wet electrode  6300  and passed it on to rectification circuitry  2919  (1.5 mm thickness, Au electroplating pad). The receiver  2903  rectifier  2919  was shunted by an L-C tank  2917  (SMD type) at the input and was itself built by the bridge configuration of 4 RF Schottky diodes (Skyworks, SMS7621-060) with one input node floating. Commercial SMD-type capacitor  2902  and resistor  2921  were used for storage and loading. The transmitter  2901  PCB was used to couple the electric field (30 MHz˜90 MHz) onto the human body  105 . The transmitter  2901  output ˜1.2 mW with 3 V pp . The L-C tank  2917  shunted across the rectifier input provided both power and voltage boosting. The resonance frequency was the same as energy delivery frequency, and the inductor and capacitor values were selected based on the L-C resonance equation f=√{square root over (¼π 2 LC)}, as well as the Q factor considerations. A load capacitor  2902  of 10 nF was shunt across the rectifier  2919  output for energy storage. Oscilloscope (Keysight, EXA Signal Analyser, N9010A) was then used for output voltage measurement, from which DC power rectified was calculated by P=V 2 /R 10ad  Two oscilloscope probes (Keysight, N2843A, 10:1) for channel 1 and channel 2 inputs were used to probe at the two nodes of the rectifier  2919  output, respectively, with both group leads connected together and left floating. The voltage difference between these two channels was taken as the output voltage in order to counteract potential fluctuations due to ambient noises. 
     The open circuit voltage was measured with an oscilloscope (Keysight, EXA Signal Analyser, N9010A) by removing all resistive components from the rectifier  2919  load at the receiver  2903  side, and shunting the oscilloscope probe (Keysight, N2843A, 10:1) across the load capacitor  2902 . The impedance of the oscilloscope probe was 10 MΩ. 
       FIG.  30    shows the correlation between the power recovered and the enhancement in transmitter  2901  output voltage/power for an on-body distance of 15 cm and  FIG.  31    shows the power recovered with respect to on-body distance for an arm to arm path with transmitter  2901  output voltage of 3V PP  ( 3103 ) and an ankle-forehead path also with transmitter  2910  output voltage of 3V PP  ( 3101 ). The recovered power positively correlated with the transmission power and negatively correlated with the increase in transmission distance. At 15 cm apart, 0.8 mW was recovered with the apparatus  2900 , when the power transmitter output 10 V PP . At the same distance, when the transmitter  2901  output reduced to 3 V PP  and ˜1.2 mW, 53 μW was recovered ( FIG.  30   ). As the distance increased, with the power transmitter  2901  placed at one wrist emitting ˜1.2 mW at 3 V pp , 1.1 μW was recovered at the other wrist. This distance was estimated to be 120 cm considering an adult&#39;s arm length (male; height 170 cm). When the transmitter  2901  was placed at the ankle and the receiver  2903  at the forehead (with the equivalent on-body transmission distance of 160 cm), 2μW was recovered (3 V PP  transmitter  2901  output) ( FIG.  31   ). The higher recovered power compared to the wrist case was attributable to the higher capacitive coupling with the Earth ground when one node is placed at the ankle. Thus, apparatus  100 ,  2900  according to embodiments may enable a large coverage range. 
     The body-coupled power transmission scheme employed was strictly below the specific absorption rate (SAR) limit for the whole body and local body parts. Taking the power transmitter  2901  to be introducing electric field with an output power of ˜1.2 mW at 20 MHz˜100 MHz, the whole-body SAR was calculated to be 2×10 −5  W/kg using the following equation and considering an average body weight of 60 kg 
       Whole body SAR≤1.2 mW/60 kg=2×10 −5  W/kg
 
     This is far below the FCC (Federal Communications Commission) limit of 1.6 W/kg. The local SAR could be calculated with a forearm section of 5 cm for estimation, with the forearm section considered as a cylindrical of 22 cm circumference with 985 kg/m 3  density (the average density of the human body): 
       Local SAR (forearm)≤1.2 mW/((π×(0.22 m/4π) 2 )×0.05 m×985 kg/m 3 )=0.025 W/kg
 
     This shows an estimated value of 0.025 W/kg, again much below the limit of 1.6 W/kg. With the enhancement in transmitter  2901  voltage to 10 V PP  (“′15 mW output power) ( FIG.  30   ), the whole-body SAR and local SAR (forearm) were calculated to be 2.5×10 −4  W/kg and 0.31 W/kg, respectively, which are below the FCC standard of 1.1 W/kg. Moreover, apart from the AC signals coupled on the body, there was no DC current flow through the human body  105  in both cases, as the return path  2927  was formed by the capacitive coupling between one node of the rectifier  2919  input and the Earth ground. 
       FIG.  32    shows the recovered power and open-circuit voltage at a receiver  2903  placed 15 cm apart from the transmitter  2701  versus the number of simultaneous receivers  2903  in the network. The transmitter  2901  output was 3 V PP  and ˜1.2 mW. Despite measurements induced fluctuations, the recovered power and open-circuit voltage were not affected by the number of other receiver  2903  nodes concurrently extracting power introduced by the body coupling, with the transmitter  2901  power of ˜1.2 mW. 
     Thus, initiating simultaneous one-to-many power transmission may be possible using apparatus  100 ,  2900  according to embodiments. 
       FIG.  33    shows the measured recovered power and open-circuit voltage versus electrode  6200 ,  6300  size of the transmitter  2901  and receiver  2903 , which were placed 15 cm apart. A similar amount of power and open circuit voltage was recovered irrespective of electrode  6300  size. Unlike the far-field RF power transmission and near-field inductive coupling where the antenna/coil size is determined by its operating frequency, therefore apparatus  100 ,  2900  according to embodiments may enable down-scaling of the sensor node and the network as a whole. 
       FIGS.  34 ,  35  and  36    show body channel gain characteristics as a power transmission medium. Probing across the receiver input, channel gain from the power transmitter  2901  to the receiver  2903  at frequencies ranging from 20 MHz to 100 MHz is illustrated;  FIG.  34    shows 120 cm wrist-to-wrist channel gain measured on three different subjects  3401 ,  3403  and  3405 ;  FIG.  35    shows channel gain along human arm measured at three different lengths: 5 cm ( 3501 ), 10 cm ( 3503 ), and 30 cm ( 3505 );  FIG.  36    shows channel gain measured along heterogeneous body paths: wrist-to-wrist ( 3605 ), wrist-to-leg ( 3603 ), and wrist-to-chest ( 3601 ). Power transmission via body coupling was found to manifest in the frequency of the megahertz range. Though the absolute amount of the pathloss was dependent on the individual, distance, delivery path, and environmental factors, 30 MHz to 90 MHz exhibited the lowest path loss, and thus optimal frequency range for body-coupled power transmission for the apparatus  2900 . 
       FIG.  37    shows the Body-coupled power received when an RF generator served as the transmitter with the power for no external source shown by  3701  and the signal  3703  indicating the power received from a source with RF generation at 4, 8, 25, 40, 50, 64, and 80 MHz.  FIG.  38    shows the corresponding results when the battery-operated PCB as described above served as the transmitter  2901 , with the power for no external source shown as signal  3801  and the signal  3803  indicating the power received the PCB source at 4, 8, 25, 40, 50, 64, and 80 MHz. This illustrates the effectiveness of the introduction of an electric field by a power transmitter  2901  where the received power across the rectifier  2919  inputs at a receiver  2903 , was at least 60 dBm higher than that without any external sources. 
       FIG.  39    shows the human body channel gain in comparison with RF approach (air-coupled; via antenna).  3901  represents the channel behaviour against increasing transmission distance when using the human body  105  as the medium and electrodes  6200 ,  6300  as the interface.  3903  represents the channel behaviour against increasing transmission distance when using the antenna for power transmission at 2.4 GHz on the body surface (results taken from Fort, A. et al. Ultra-wideband channel model for communication around the human body. IEEE I. Sel. Areas Commun. 24, 927-933 (2006)). Thus, unlike conventional far-field RF power transmission approach which may be limited by line of sight (LoS) requirements due to body-shadowing effects, the body-coupled power recovery may be insusceptible to body shadowing effect and may achieve around 30˜70 dB higher received power as compared to RF transmission at 2.4 GHz on/around the body and up to 50 dB higher compared to RF transmission at 900 MHz. 
       FIGS.  40 - 43    show Improvements in power recovery with the use of detuned L-C boosting at the power receiver  2903 . Specifically,  FIG.  40    shows the recovered power with and without the impedance/voltage boosting techniques using the L-C tank  2917  (at 5 cm apart at 4 MHz, 8 MHz, 25 MHz, 40 MHz, 50 MHz, 64 MHz, and 80 MHz).  FIG.  41    shows the open circuit voltage with and without the impedance/voltage boosting techniques using the L-C tank  2917  (likewise, the receiving node is placed 5 cm apart).  FIG.  42    shows the improvement in power recovery (illustrated as boosting factor, i.e. the recovered power with the L-C tank  2917  and that without the L-C tank  2917 ) at the power transmission frequency of 25 MHz ( 4301 ), 64 MHz ( 4303 ), and 80 MHz ( 4305 ). The resistive load  2921  across the receiver  2903  output was 10 kΩ.  FIG.  43    shows power recovery along an on-body path from one wrist to the other wrist with the L-C boosting technique, in comparison to no L-C boosting being employed. The distance of 120 cm corresponds to the subject&#39;s arm length, i.e. from one wrist to the other wrist. When attempting for power recovery without L-C boosting, the successful power recovery occurred only at 15 cm or below, which roughly corresponds to from wrist to elbow. 
     Thus, with the impedance boosting techniques according embodiments, up to 13 times more power was recovered at the receiver  2903 , and the body area coverage was enhanced to the entire arm length (120 cm). 
       FIGS.  44  to  46    show the impedance for maximum power extraction with the receiver  2903  produced as described above, specifically with the recovered power with respect to load resistance at the power transmission frequency of: 40 MHz ( FIG.  44   ); 64 MHz ( FIG.  45   ); and 80 MHz ( FIG.  46   ). The power transmitter outputted ˜1.2 mW with 3 V PP , and was placed 5 cm apart from the receiver. The maximum power extraction was observed to occur at the load impedance of 10 kΩ. 
     In order to characterise the usefulness of the preferred embodiment in the power harvesting mode, a receiver  2903  according to the second embodiment received the signal via a wet electrode  6300  and passed it on to rectification circuitry  2919  (1.5 mm thickness, Au electroplating pad). The receiver (harvester) rectifier  2919  was built by the bridge configuration of 4 RF PIN diodes (Infineon, BAR6303WE6327HTSA 1 ). Commercial SMD-type capacitor  2902  and resistor  2921  were used for storage and loading. A wet electrode  300  (Red dot, 3M Ag/AgCl,  2237 ) was attached to the skin to transfer the charges coupled on the body onto one input node of the receiver  2903 . Portable picoscope (Pico Technology,  4424 ) were used for output voltage measurement, from which DC power rectified was calculated by P=V store   2 /R load . A picoscope probe (Pico Technology, 10:1) was used to probe at the rectifier  2919  output to determine V store . An analogue buffer (Analog Devices, AD8065) with a common mode impedance of 1000 GΩ at 2.1 pF was inserted in between the resistive load and the picoscope probes, in order to eliminate the loading effect caused by the 10 MΩ probes. 
       FIG.  47    shows the power harvested with a 30 MΩ load resistance under 7 environmental settings. #1 and #2 were measured under different office settings; #3 was measured inside a café; #4 and #5 were measured inside an apartment; #6, #7, and #8 were measured inside different libraries/studies. In order to help ensure a fair representation of power recoverability at all potential locations for the particular environment, 30 measurements were performed at various locations and with different body positioning under each environmental setting. The measurements show that around 20 nW˜2.2 μW may be recovered, meaning that receivers  301 ,  2903  according to embodiments may be able to continuously scavenge the body-coupled ambient EM waves in all environments, which may be helpful in the absence of an active power transmitting source. 
       FIG.  48    shows the power harvested with a 30 MΩ load resistor by 5 receivers  2903  placed on different parts of the body simultaneously, under environment #2. Power harvested from 3 individuals is illustrated. Due to individual-based differences in tissue electrical characteristics, ambient energy recoverable varies from person to person even with the same posture and positioning under the same environment. However, as can be seen from  FIG.  48   , for one individual, the amount of power harvested may remain similar regardless of where the electrodes  6300  were placed, despite slight fluctuations. Among the locations are those with little movement/pressure-induced electrostatics and little daylight coverage due to clothing. Such homogeneity in terms of power harvested at various placements indicates the potential of a wide body area coverage with the devices according to embodiments, which may allow devices placed on chest such as electrocardiogram (ECG) sensors  301   e  to operate on the sustainable power supply. Moreover, with multiple receivers  301   a ,  301   b ,  301   c ,  301   d ,  301   e ,  301   f  operating simultaneously, the amount of power recovered may remain similar to the amount harvested by one single receiver  301 , 2903 , which may indicate the potential for multi-channel energy harvesting without power degradation in devices according to embodiments. 
       FIG.  49    shows the power harvested with a 30 MΩ resistor and the open-circuit voltage achieved with various electrode sizes. The resistor  2903  was placed on a subject&#39;s right wrist and measured under environment #2. With the reduction of electrode  6300  size, the amount of power remained similar with fluctuations, suggesting that electrode  300 ,  6300  size may be of little relevance in power recoverability using receiver  301 ,  2903  according to embodiments and that they may have the potential for device downscaling.  FIG.  50    shows the power harvested from 4 different environmental scenarios (an office  5001 , a café  5003 , a first home  5005  and a second home  5007 ) against the load resistance of 1 MΩ to 50 MΩ, where the resistance of around 30 MΩ was observed to be the optimal power extraction point for ambient energy harvesting. Note that this was significantly different from the 10 kΩ observed to be the optimal power extraction point for the receiver  2903  operating with a power transmitter  2901  ( FIGS.  44  to  46   ). Thus, apparatus  100  according to the preferred embodiment which may achieve a very wide range of load resistance by varying T 0   1703  in the dual-mode buck-boost converter  309  may be effectively implemented in both harvesting and power transfer modes. 
       FIG.  51    illustrates schematically an energy transfer apparatus  5100  according to a third embodiment for powering wearable devices via energy transfer via the body and energy harvesting. In line with the above described embodiments, the apparatus comprises receivers  5102  and a transmitter  5101  coupled to the human body. 
     Each receiver  5102  includes a matching network  5103  which, in the active mode, is used to maximize the power transfer efficiency from transmitter  5101  to receiver  5102 . The matching may be modified depending on the human subject by changing variable C. Each receiver  5102  further includes a full-bridge rectifier  5105  with a store capacitor  5123  in which body-coupled ambient energy  101  and the active transferred energy from power TX  5101  are changed into DC voltage. Considering the energy central frequency and energy efficiency, a diode in the full-bridge rectifier  5105  may be an active switch based on the ZVS or ZCS technique. The receiver  5102  further comprises a DC-DC converter  5017  which scales voltage into supply voltage for use rectified voltage in a wearable device. The DC-DC converter  5107  is suitable for use due to its high-power efficiency and up/down voltage conversion, and further includes a low drop out regulator (LDO; not shown) which supports low-power capability and only down conversion. A power amplifier  5109 , equivalently receiver controller, senses the output voltage of the full-bridge rectifier  5105  and sends the information to the Power TX  5101  to achieve maximum power efficiency. It also generates the control signal for the DC-DC converter  5107  and changes the matching network  5103  parameters depending on the situation. A data transmitter  5111  uses a body area network (BAN) for data transmission for its availability using a transmitter  5111  on the millimetre-scale. Different frequencies are employed for power and data communication to prevent confusion. 
     The transmitter  5101  includes a BAN data receiver  5113  which can use the information and mode received from the power receiver  5102 , such as rectifier  5105  output voltage to transmit better power efficiency. For the data acquisition from the receiver  5102 , the BAN receiver  5113  receives data using a body area network (BAN). A digitally controlled, voltage-controlled oscillator (VCO)  5115  determines active energy central frequency. For low-power consumption, the VCO  5115  uses high threshold MOSFET for leakage current reduction. A power amplifier  5117  (PA) generates active body-coupled energy pulses with a high power-efficient Class D or E amplifier using the output pulse of the VCO  5115 . A matching network  5119  is employed to boost power efficiency. A transmitter controller  5121  controls all sub-blocks of the Power TX  5101 . Based on the data from the RX  5102  via BAN, the Controller  5121  determines all of the parameters and generates a control signal such as frequency and a maximum current of the PA  5117  for safety. A current limiter  5122  sets the maximum current consumed in the PA  5117 , which limits the maximum power transfer. A binary-weighted current mirror is used to control the current level digitally. 
     Having now fully described the invention, it should be apparent to one of ordinary skill in the art that many modifications can be made hereto without departing from the scope as claimed, including the combining of individual functional components or modules of the different embodiments described.