Patent Publication Number: US-2021167695-A1

Title: Power converting apparatus, motor driving apparatus, and air conditioner

Description:
FIELD 
     The present invention relates to a power converting apparatus that converts an alternating-current power supplied from an alternating-current power supply into a direct-current power, and a motor driving apparatus and an air conditioner that include the power converting apparatus. 
     BACKGROUND 
     A power supply current that is a current supplied from a power supply includes a harmonic current. The harmonic current is a frequency component with a frequency higher than the frequency of a fundamental wave. In order to reduce failures caused by a harmonic current, international restrictions are imposed on electronic devices generating harmonic currents. In compliance with the restrictions, measures for reducing harmonic currents included in power supply currents by chopping of an alternating current (AC) or a direct current (DC) are taken in converters. 
     Among such converters, bridgeless converters in which a rectifier circuit is constituted by switching elements have been actively examined as a technology for reducing losses by using the AC chopping technology. A direct-current power supply device, which is an example of the bridgeless converters, described in Patent Literature 1 includes a first arm constituted by an upper diode and a lower diode connected in series to each other, a second arm constituted by an upper switching element and a lower switching element connected in series to each other, and a direct-current power supply for driving the second arm. The direct-current power supply device described in Patent Literature 1 also includes a first drive circuit that uses a voltage output from the direct-current power supply as a power supply voltage to generate a driving signal for driving the lower switching element of the second arm, a bootstrap circuit that uses the voltage output from the direct-current power supply to generate a voltage for driving the upper switching element of the second arm, and a second drive circuit that uses the voltage output from the bootstrap circuit as a power supply voltage to generate a driving signal for driving the upper switching element of the second arm. Hereinafter, the drive circuits will be referred to as driving circuits. In addition, hereinafter, the upper switching element of the second arm will be simply referred to as an upper switching element, and the lower switching element of the second arm will be simply referred to as a lower switching element. 
     The bootstrap circuit is constituted by a resistor, a diode, and a capacitor. In the technology described in Patent Literature 1, when the lower switching element is turned ON, a closed circuit is formed by the direct-current power supply, the bootstrap circuit, and the lower switching element, and the capacitor of the bootstrap circuit is thus charged by the direct-current power supply. In this process, in addition to the voltage of the direct-current power supply, a forward voltage of the body diode formed in the lower switching element of the second arm is also applied to the capacitor. The capacitor voltage of the charged capacitor is then used as the power supply voltage for the second driving circuit, and a driving signal for driving the upper switching element is thus generated in the second driving circuit. 
     CITATION LIST 
     Patent Literature 
     Patent Literature 1: Japanese Patent Application Laid-open No. 2016-220378 
     SUMMARY 
     Technical Problem 
     In a case where metal-oxide-semiconductor field-effect transistors (MOSFETs) made of wide band gap (WBG) semiconductors, for example, are used for the switching elements, a potential barrier of a p-n junction of a WBG semiconductor is higher than that of a silicon (Si) semiconductor. Thus, a voltage at which a forward current starts to flow in a body diode formed in a WBG MOSFET is a value higher than a voltage at which a forward current starts to flow in a body diode formed in a Si switching element. It can thus be said that the forward current-forward voltage characteristics of a body diode formed in a WBG MOSFET is inferior to the forward current-forward voltage characteristics of a body diode formed in a Si switching element. In a case where a switching element in which a voltage at which a forward current starts to flows in a body diode is relatively high as described above is used for a lower switching element of Patent Literature 1, the capacitor voltage of the capacitor of the bootstrap circuit, that is, the power supply voltage for the driving circuits may be higher than a rated voltage of a driving circuit. When a power supply voltage higher than the rated voltage of a driving circuit is applied to the driving circuit in this manner, there is a problem in that a withstand voltage of the driving circuit decreases. The withstand voltage used herein is a voltage that can be applied to a driving circuit for a prescribed time without causing breakdown of the driving circuit. In addition, because the value of a driving signal generated by a driving circuit becomes larger as the power supply voltage for the driving circuit is higher, there is a problem in that a short circuit withstand of the upper switching element decreases. The short circuit withstand is defined as a time from when a short-circuit current starts to flow into the upper switching element until the upper switching element is damaged. 
     The present invention has been made in view of the above, and an object thereof is to provide a power converting apparatus capable of improving the reliability by preventing or reducing an increase in a power supply voltage for driving circuits for switching elements. 
     Solution to Problem 
     To solve the aforementioned problems and achieve the object, a power converting apparatus according to the present invention is a power converting apparatus for converting an alternating-current power supplied from an alternating-current power supply into a direct-current power, and includes: a first line and a second line, each of the first line and the second line being connected to the alternating-current power supply; and a first reactor disposed on the first line. The power converting apparatus includes: a first arm including a first switching element, a second switching element, and a third line having a first connection point, the first switching element being connected to the second switching element in series by the third line, the first connection point being connected to the first reactor by the first line. The power converting apparatus includes: a second arm connected in parallel with the first arm and including a third switching element, a fourth switching element, and a fourth line having a second connection point, the third switching element being connected to the fourth switching element in series by the fourth line, the second connection point being connected to the alternating-current power supply by the second line. The power converting apparatus includes: a first capacitor connected in parallel with the second arm; a first driving circuit outputting a first driving signal for driving the first switching element; a bootstrap circuit including a second capacitor, the second capacitor applying a power supply voltage for the first driving circuit to the first driving circuit; and a diode adjusting the power supply voltage, wherein a first voltage is lower than a second voltage, the first voltage being a voltage at which a forward current starts to flow in the diode, the second voltage being a voltage at which a forward current starts to flow in a body diode formed in the second switching element. 
     Advantageous Effects of Invention 
     The power converting apparatus according to the present invention produces an effect of being capable of improving the reliability by preventing or reducing an increase in the power supply voltage for the driving circuits for the switching elements. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a diagram illustrating an example of a configuration of a power converting apparatus according to a first embodiment. 
         FIG. 2  is a schematic cross-sectional view illustrating an outline structure of a MOSFET that can be used as switching elements illustrated in  FIG. 1 . 
         FIG. 3  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of a power supply current is larger than a current threshold and a power supply voltage polarity is positive. 
         FIG. 4  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is negative. 
         FIG. 5  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is positive. 
         FIG. 6  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is negative. 
         FIG. 7  is a first diagram for explaining an operation that causes a capacitor short circuit via an alternating-current power supply and a reactor in the power converting apparatus according to the first embodiment. 
         FIG. 8  is a second diagram for explaining an operation that causes a capacitor short circuit via the alternating-current power supply and the reactor in the power converting apparatus according to the first embodiment. 
         FIG. 9  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is positive. 
         FIG. 10  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is negative. 
         FIG. 11  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is positive. 
         FIG. 12  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is negative. 
         FIG. 13  is a diagram illustrating an example of a configuration of a control unit of the power converting apparatus according to the first embodiment. 
         FIG. 14  is a chart illustrating an example of a power supply voltage, and a power supply voltage phase estimation value and a sinusoidal value calculated by a power supply voltage phase calculating unit illustrated in  FIG. 13 . 
         FIG. 15  is a diagram illustrating an example of a configuration of a first pulse generating unit of the power converting apparatus according to the first embodiment. 
         FIG. 16  is a chart illustrating an example of a reference ON-duty, a carrier wave, and a reference pulse width modulation (PWM) signal in  FIG. 15 . 
         FIG. 17  is a chart illustrating an example of the reference PWM signal, an inverted PWM signal, a first PWM signal, and a second PWM signal in  FIG. 15 . 
         FIG. 18  is a flowchart illustrating an example of procedures of a selecting process performed by a pulse selector of the first pulse generating unit illustrated in  FIG. 15 . 
         FIG. 19  is a schematic graph illustrating the relation of currents flowing through a switching element and a body diode illustrated in  FIG. 1 , the loss of the switching element, and the loss of the body diode. 
         FIG. 20  is a flowchart illustrating an example of procedures of a process performed by a second pulse generating unit illustrated in  FIG. 13 . 
         FIG. 21  is a flowchart illustrating an example of procedures for controlling the switching elements on the basis of the power supply current by the second pulse generating unit illustrated in  FIG. 13 . 
         FIG. 22  is a chart illustrating a first example of signals, corresponding to one cycle of the power supply voltage, generated in the power converting apparatus according to the first embodiment. 
         FIG. 23  is a chart illustrating a second example of signals, corresponding to one cycle of the power supply voltage, generated in the power converting apparatus according to the first embodiment. 
         FIG. 24  is a chart illustrating an example of signals when the power converting apparatus according to the first embodiment performs simple switching control. 
         FIG. 25  is a chart illustrating an example of signals in passive states generated by the power converting apparatus according to the first embodiment. 
         FIG. 26  is a diagram illustrating driving circuits and bootstrap circuits included in the power converting apparatus according to the first embodiment. 
         FIG. 27  is a diagram illustrating an example of a configuration of a power converting apparatus according to a first modification of the first embodiment. 
         FIG. 28  is a diagram illustrating an example of a configuration of a power converting apparatus according to a second modification of the first embodiment. 
         FIG. 29  is a diagram illustrating an example of a configuration of a power converting apparatus according to a third modification of the first embodiment. 
         FIG. 30  is a diagram illustrating an example of a configuration of a power converting apparatus according to a second embodiment. 
         FIG. 31  is a diagram illustrating an example of a hardware configuration implementing the control unit of the first and second embodiments. 
         FIG. 32  is a diagram illustrating an example of a configuration of a motor driving apparatus according to a third embodiment. 
         FIG. 33  is a diagram illustrating an example of a configuration of an air conditioner according to a fourth embodiment. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     A power converting apparatus, a motor driving apparatus, and an air conditioner according to certain embodiments of the present invention will be described in detail below with reference to the drawings. Note that the present invention is not limited to the embodiments. 
     First Embodiment 
       FIG. 1  is a diagram illustrating an example of a configuration of a power converting apparatus according to a first embodiment. A power converting apparatus  100  according to the first embodiment is a power supply device having an AC-DC converting function for converting an alternating-current power supplied form a single-phase alternating-current power supply  1  into a direct-current power and applying the direct-current power to a load  50 . Hereinafter, the single-phase alternating-current power supply  1  may simply be referred to as an alternating-current power supply  1 . As illustrated in  FIG. 1 , the power converting apparatus  100  includes a reactor  2 , which is a first reactor, a bridge circuit  3 , a smoothing capacitor  4 , which is a first capacitor, a power supply voltage detecting unit  5 , a power supply current detecting unit  6 , a bus voltage detecting unit  7 , and a control unit  10 . 
     The bridge circuit  3  includes a first arm  31 , which is a first circuit, and a second arm  32 , which is a second circuit. The first arm  31  includes a switching element  311  and a switching element  312 , which are connected in series. A body diode  311   a  is formed in the switching element  311 . The body diode  311   a  is connected in parallel between a drain and a source of the switching element  311 . A body diode  312   a  is formed in the switching element  312 . The body diode  312   a  is connected in parallel between a drain and a source of the switching element  312 . The body diodes  311   a  and  312   a  are each used as a freewheeling diode. 
     The second arm  32  incudes a switching element  321  and a switching element  322 , which are connected in series. The second arm  32  is connected in parallel with the first arm  31 . A body diode  321   a  is formed in the switching element  321 . The body diode  321   a  is connected in parallel between a drain and a source of the switching element  321 . A body diode  322   a  is formed in the switching element  322 . The body diode  322   a  is connected in parallel between a drain and a source of the switching element  322 . The body diodes  321   a  and  322   a  are each used as a freewheeling diode. 
     Specifically, the power converting apparatus  100  incudes a first line  501  and a second line  502 , which are each connected to the alternating-current power supply  1 , and the reactor  2  disposed on the first line  501 . In addition, the first arm  31  includes the switching element  311 , which is a first switching element, the switching element  312 , which is a second switching element, and a third line  503  having a first connection point  506 . The switching element  311  is connected in series to the switching element  312  by the third line  503 . The first line  501  is connected to the first connection point  506 . The first connection point  506  is connected to the alternating-current power supply  1  via the first line  501  and the reactor  2 . 
     The second arm  32  includes the switching element  321 , which is a third switching element, the switching element  322 , which is a fourth switching element, and a fourth line  504  having a second connection point  508 . The switching element  321  is connected in series to the switching element  322  by the fourth line  504 . The second line  502  is connected to the second connection point  508 . The second connection point  508  is connected to the alternating-current power supply  1  via the second line  502 . The smoothing capacitor  4 , which is a capacitor, is connected in parallel with the second arm  32 . 
     MOSFETs formed of WBG semiconductors can be used for the switching elements  311 ,  312 ,  321 , and  322 . For the WBG semiconductors, gallium nitride (GaN) materials, silicon carbide (SiC), diamond, or aluminum nitride is used. Use of the WBG semiconductors for the switching elements  311 ,  312 ,  321 , and  322  increases the withstand voltage characteristics and also increases the allowable current density, which allows miniaturization of modules. In addition, because the WBG semiconductors have high heat resistance, use of the WBG semiconductors for the switching elements  311 ,  312 ,  321 , and  322  allows miniaturization of radiating fins for radiating heat generated by the switching elements. 
     The control unit  10  generates driving pulses for causing the switching elements  311 ,  312 ,  321 , and  322  of the bridge circuit  3  to operate on the basis of signals output from the power supply voltage detecting unit  5 , the power supply current detecting unit  6 , and the bus voltage detecting unit  7 . The power supply voltage detecting unit  5  detects a power supply voltage Vs, which is a voltage output from the alternating-current power supply  1 , and outputs an electrical signal indicating the detection result to the control unit  10 . The power supply current detecting unit  6  detects a power supply current Is, which is a current output from the alternating-current power supply  1 , and outputs an electrical signal indicating the detection result to the control unit  10 . The bus voltage detecting unit  7  detects a bus voltage Vdc, and outputs the detected bus voltage Vdc to the control unit  10 . The bus voltage Vdc is a voltage obtained by smoothing a voltage output from the bridge circuit  3  by the smoothing capacitor  4 . 
     Next, basic operation of the power converting apparatus  100  according to the first embodiment will be described. Hereinafter, the switching elements  311  and  321  connected to the positive side of the alternating-current power supply  1 , that is, a positive terminal of the alternating-current power supply  1  may also be referred to as upper switching elements. In addition, the switching elements  312  and  322  connected to the negative side of the alternating-current power supply  1 , that is, a negative terminal of the alternating-current power supply  1  may also be referred to as lower switching elements. 
     In the first arm  31 , the upper switching element and the lower switching element operate complementarily. Specifically, when one of the upper switching element and the lower switching element is ON, the other is OFF. The switching elements  311  and  312  constituting the first arm  31  are driven by driving signals output from driving circuits, which will be described later. The driving circuits amplify PWM signals generated by the control unit  10 , and output the amplified signals as driving signals. The operations of turning the switching elements ON or OFF in accordance with driving signals will hereinafter also be referred to as switching operations. 
     The switching elements  321  and  322  constituting the second arm  32  perform operations in accordance with driving signals to be turned ON or OFF in a manner similar to the switching elements  311  and  312 . Basically, the switching elements are turned ON or OFF depending on a power supply voltage polarity that is the polarity of voltage output from the alternating-current power supply  1 . Specifically, when the power supply voltage polarity is positive, the switching element  322  is ON and the switching element  321  is OFF, and when the power supply voltage polarity is negative, the switching element  321  is ON and the switching element  322  is OFF. In the first embodiment, however, as will be described later, in order to prevent a short circuit of the smoothing capacitor  4  via the alternating-current power supply  1  and the reactor  2 , the switching element  322  and the switching element  321  are both OFF when the absolute value of the power supply current Is output from the alternating-current power supply  1  is equal to or smaller than a threshold. Alternatively, in order to prevent a short circuit of the smoothing capacitor  4  via the alternating-current power supply  1  and the reactor  2 , the switching element  312  and the switching element  311  may both be OFF when the absolute value of the power supply current Is output from the alternating-current power supply  1  is equal to or smaller than a threshold. Hereinafter, the threshold to be compared with the absolute value of the power supply current Is will be referred to as a current threshold. In addition, hereinafter, the short circuit of the smoothing capacitor  4  will be referred to as a capacitor short circuit. The capacitor short circuit is a state in which the energy stored in the smoothing capacitor  4  is released and the current is regenerated back to the alternating-current power supply  1 . 
     Next, the relation between the states of the switching elements in the first embodiment and the path of current flowing in the power converting apparatus  100  according to the first embodiment will be explained. Note that the structure of the MOSFETs will be described with reference to  FIG. 2  before the explanation. 
       FIG. 2  is a schematic cross-sectional view illustrating an outline structure of a MOSFET that can be used as the switching elements illustrated in  FIG. 1 .  FIG. 2  illustrates an n-type MOSFET as an example. In an n-type MOSFET, a p-type semiconductor substrate  600  is used as illustrated in  FIG. 2 . A source electrode S, a drain electrode D, and a gate electrode G are formed on the semiconductor substrate  600 . High-concentration impurity is introduced by ion implantation into portions in contact with the source electrode S and the drain electrode D to form n-type regions  601 . In addition, an insulating oxide layer  602  is formed between a portion of the semiconductor substrate  600  where no n-type region  601  is formed and the gate electrode G. Thus, the insulating oxide layer  602  is present between the gate electrode G and a p-type region  603  of the semiconductor substrate  600 . 
     When a positive voltage is applied to the gate electrode G, electrons are attracted to an interface between the p-type region  603  and the insulating oxide layer  602  of the semiconductor substrate  600 , and the interface is negatively charged. The electron density of a portion where electrons have gathered becomes higher than a hole density, and the portion becomes n-type. The portion that has become n-type functions as a current path, and will be referred to as a channel  604 . The channel  604  is an n-type channel in the example of  FIG. 2 . When the MOSFET is controlled to be ON, more current flows to the channel  604  than to a body diode formed in the p-type region  603 . 
       FIGS. 3 to 6  illustrate current paths in the power converting apparatus  100  according to the first embodiment when the absolute value of the power supply current Is is larger than the current threshold. 
       FIG. 3  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is positive. In  FIG. 3 , the power supply voltage polarity is positive, the switching element  311  and the switching element  322  are ON, and the switching element  312  and the switching element  321  are OFF. In this state, current flows in the order of the alternating-current power supply  1 , the reactor  2 , the switching element  311 , the smoothing capacitor  4 , the switching element  322 , and the alternating-current power supply  1 . Thus, in the first embodiment, a synchronous rectification operation is performed in such a manner that current flows through each of the channels of the switching element  311  and the switching element  322  instead of flowing through the body diode  311   a  and the body diode  322   a.    
       FIG. 4  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is negative. In  FIG. 4 , the power supply voltage polarity is negative, the switching element  312  and the switching element  321  are ON, and the switching element  311  and the switching element  322  are OFF. In this state, current flows in the order of the alternating-current power supply  1 , the switching element  321 , the smoothing capacitor  4 , the switching element  312 , the reactor  2 , and the alternating-current power supply  1 . Thus, in the first embodiment, synchronous rectification operation is performed in such a manner that current flows through each of the channels of the switching element  321  and the switching element  312  instead of flowing through the body diode  321   a  and the body diode  312   a.    
       FIG. 5  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is positive. In  FIG. 5 , the power supply voltage polarity is positive, the switching element  312  and the switching element  322  are ON, and the switching element  311  and the switching element  321  are OFF. In this state, current flows in the order of the alternating-current power supply  1 , the reactor  2 , the switching element  312 , the switching element  322 , and the alternating-current power supply  1 , and a power supply short-circuit path that does not pass through the smoothing capacitor  4  is thus formed. Thus, in the first embodiment, the power supply short-circuit path is formed in such a manner that current flows through each of the channels of the switching element  312  and the switching element  322  instead of flowing through the body diode  312   a  and the body diode  322   a.    
       FIG. 6  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is larger than the current threshold and the power supply voltage polarity is negative. In  FIG. 6 , the power supply voltage polarity is negative, the switching element  311  and the switching element  321  are ON, and the switching element  312  and the switching element  322  are OFF. In this state, current flows in the order of the alternating-current power supply  1 , the switching element  321 , the switching element  311 , the reactor  2 , and the alternating-current power supply  1 , and a power supply short-circuit path that does not pass through the smoothing capacitor  4  is formed. Thus, in the first embodiment, the power supply short-circuit path is formed in such a manner that current flows through each of the channels of the switching element  311  and the switching element  321  instead of flowing through the body diode  311   a  and the body diode  321   a.    
     The control unit  10  can control the values of the power supply current Is and the bus voltage Vdc by controlling switching of the current paths described above. 
     When the switching element  311  and the switching element  322  are turned ON while the power supply current Is is not flowing, however, a capacitor short circuit via the alternating-current power supply  1  and the reactor  2  occurs. As a result, current flows in a direction opposite to the normal direction, which may cause such problems as degradation in power factor, increase in harmonic components, damage to an element due to overcurrent, or increase in loss. 
       FIGS. 7 and 8  illustrate states in which a capacitor short circuit via the alternating-current power supply  1  and the reactor  2  occurs. 
       FIG. 7  is a first diagram for explaining an operation that causes a capacitor short circuit via the alternating-current power supply and the reactor in the power converting apparatus according to the first embodiment.  FIG. 7  illustrates a state in which the power supply voltage polarity is positive, and the power supply current Is does not flow. Because the power supply voltage polarity is positive, current should normally flow in the order of the alternating-current power supply  1 , the reactor  2 , the switching element  311 , the smoothing capacitor  4 , the switching element  322 , and the alternating-current power supply  1  as illustrated in  FIG. 3 . When the switching element  311  and the switching element  322  are turned ON while the power supply current Is is not flowing, however, current flows in the direction opposite to the normal direction and a capacitor short circuit thus occurs as illustrated in  FIG. 7 . Thus, the energy stored in the smoothing capacitor  4  is output to the alternating-current power supply  1 . 
       FIG. 8  is a second diagram for explaining an operation that causes a capacitor short circuit via the alternating-current power supply and the reactor in the power converting apparatus according to the first embodiment.  FIG. 8  illustrates a state in which the power supply voltage polarity is negative, and the power supply current Is does not flow. Because the power supply voltage polarity is negative, current should normally flow in the order of the alternating-current power supply  1 , the switching element  321 , the smoothing capacitor  4 , the switching element  312 , the reactor  2 , and the alternating-current power supply  1  as illustrated in  FIG. 4 . When the switching element  312  and the switching element  321  are turned ON in the case where the power supply current Is is not flowing, however, current flows in the direction opposite to the normal direction and a capacitor short circuit occurs as illustrated in  FIG. 8 . 
     In order to prevent a capacitor short circuit, the power converting apparatus  100  according to the first embodiment permits the switching elements  321  and  322  to be in the ON state when the absolute value of the power supply current Is is equal to or larger than the current threshold, and turns the switching elements  321  and  322  OFF when the absolute value of the power supply current Is is smaller than the threshold. This enables prevention of a capacitor short circuit via the alternating-current power supply  1  and the reactor  2 , and can achieve a highly reliable power converting apparatus. 
       FIGS. 9 to 12  illustrate current paths in the power converting apparatus  100  according to the first embodiment when the absolute value of the power supply current Is is smaller than the current threshold. 
       FIG. 9  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is positive. In FIG.  9 , the power supply voltage polarity is positive, the switching element  311  is ON, and the switching element  312 , the switching element  321 , and the switching element  322  are OFF. In this case, the body diode  322   a  of the switching element  322  functions as a freewheeling diode, and current flows in the order of the alternating-current power supply  1 , the reactor  2 , the switching element  311 , the smoothing capacitor  4 , the body diode  322   a , and the alternating-current power supply  1  as illustrated in  FIG. 9 . Note that it is sufficient if the absolute value of the power supply current Is is such a value that does not cause malfunctions, and as the absolute value is smaller, the synchronous rectification period is longer and conduction loss can be reduced more effectively. In addition, when the absolute value of the power supply current Is is such a small value that does not require the synchronous rectification operation, the switching element  311  may be turned OFF. When the switching element  311  is turned OFF, no gate driving power of the switching element  311  is generated, which can reduce power consumption for generating driving signals as compared with a case where the synchronous rectification operation is performed. Note that details of the driving circuits that generate driving signals will be described later. 
       FIG. 10  is a first diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is negative. In  FIG. 10 , the power supply voltage polarity is negative, the switching element  312  is ON, and the switching element  311 , the switching element  321 , and the switching element  322  are OFF. In this case, the body diode  321   a  of the switching element  321  functions as a freewheeling diode, and current flows in the order of the alternating-current power supply  1 , the body diode  321   a , the smoothing capacitor  4 , the switching element  312 , the reactor  2 , and the alternating-current power supply  1  as illustrated in  FIG. 10 . Note that it is sufficient if the absolute value of the power supply current Is is such a value that does not cause malfunctions, and as the absolute value is smaller, the synchronous rectification period is longer and conduction loss can be reduced more effectively. In addition, when the absolute value of the power supply current Is is such a small value that does not require the synchronous rectification operation, the switching element  312  may be turned OFF. When the switching element  312  is turned OFF, no gate driving power of the switching element  312  is generated, which can reduce power consumption for generating driving signals as compared with a case where the synchronous rectification operation is performed. 
       FIG. 11  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is positive. In  FIG. 11 , the power supply voltage polarity is positive, the switching element  312  is ON, and the switching element  311 , the switching element  321 , and the switching element  322  are OFF. In this case, the body diode  322   a  of the switching element  322  functions as a freewheeling diode, and current flows in the order of the alternating-current power supply  1 , the reactor  2 , the switching element  312 , the body diode  322   a , and the alternating-current power supply  1  as illustrated in  FIG. 11 . Note that, in this case, because a short-circuit current flows, even when the absolute value of the power supply current Is is smaller than the current threshold, the switching element  322  may be turned ON at the same time when the switching element  312  is turned ON. In this case, because a drop voltage due to an ON-resistance of the switching element  322  is smaller than a forward voltage of the body diode  322   a , the conduction loss at the switching element  322  is reduced. 
       FIG. 12  is a second diagram illustrating a path of current flowing in the power converting apparatus according to the first embodiment when the absolute value of the power supply current is smaller than the current threshold and the power supply voltage polarity is negative. In  FIG. 12 , the power supply voltage polarity is negative, the switching element  311  is ON, and the switching element  312 , the switching element  321 , and the switching element  322  are OFF. In this case, the body diode  321   a  of the switching element  321  functions as a freewheeling diode, and current flows in the order of the alternating-current power supply  1 , the body diode  321   a , the switching element  311 , the reactor  2 , and the alternating-current power supply  1  as illustrated in  FIG. 12 . Note that, in this case, because a short-circuit current flows, even when the absolute value of the power supply current Is is smaller than the current threshold, the switching element  321  may be turned ON at the same time when the switching element  311  is turned ON. In this case, because a drop voltage due to an ON-resistance of the switching element  321  is smaller than a forward voltage of the body diode  321   a , the conduction loss at the switching element  321  is reduced. 
     Next, a configuration of the control unit  10  of the power converting apparatus  100  according to the first embodiment will be described.  FIG. 13  is a diagram illustrating an example of the configuration of the control unit of the power converting apparatus according to the first embodiment. As illustrated in  FIG. 13 , the control unit  10  includes a power supply current command value control unit  21 , an ON-duty control unit  22 , a power supply voltage phase calculating unit  23 , a first pulse generating unit  24 , a second pulse generating unit  25 , a current command value calculating unit  26 , and an instantaneous value command value calculating unit  27 . 
     The power supply current command value control unit  21  calculates an effective current value command value Is_rms* from a bus voltage Vdc detected by the bus voltage detecting unit  7  and a bus voltage command value Vdc*. The bus voltage command value Vdc* may be set in advance or may be input from outside of the power converting apparatus  100 . The power supply current command value control unit  21  calculates the effective current value command value Is_rms* by proportional-integral control based on a difference between the bus voltage Vdc and the bus voltage command value Vdc*. 
     The current command value calculating unit  26  converts the effective current value command value Is_rms* into a sinusoidal command value, and outputs the sinusoidal command value. The instantaneous value command value calculating unit  27  calculates a power supply current instantaneous value command value Is* by using the effective current value command value Is_rms* calculated by the current command value calculating unit  26 , and a sinusoidal value sin θ{circumflex over ( )} s  calculated by the power supply voltage phase calculating unit  23 . 
     The ON-duty control unit  22  performs proportional-integral control on a deviation between the power supply current instantaneous value command value Is* calculated by the instantaneous value command value calculating unit  27  and the power supply current Is detected by the power supply current detecting unit  6  to calculate a reference ON-duty duty of the switching elements  311  and  312 . 
     The power supply voltage phase calculating unit  23  calculates a power supply voltage phase estimation value θ{circumflex over ( )} s  and the sinusoidal value sin θ{circumflex over ( )} s  by using the power supply voltage Vs detected by the power supply voltage detecting unit  5 .  FIG. 14  is a chart illustrating an example of the power supply voltage, and the power supply voltage phase estimation value and the sinusoidal value calculated by the power supply voltage phase calculating unit illustrated in  FIG. 13 .  FIG. 14  illustrates the power supply voltage Vs, the power supply voltage phase estimation value θ{circumflex over ( )} s , and the sinusoidal value sin θ{circumflex over ( )} s  in this order from the top. 
     The power supply voltage phase calculating unit  23  linearly increases the power supply voltage phase estimation value θ{circumflex over ( )} s , detects a timing at which the power supply voltage Vs changes from the negative polarity to the positive polarity, and resets the power supply voltage phase estimation value θ{circumflex over ( )} s  to 0 at the timing. As a result, under an ideal condition with no control delay and no detection delay, the power supply voltage phase estimation value θ{circumflex over ( )} s  becomes 360°, that is, 0° at the timing when the power supply voltage Vs is switched from the negative polarity to the positive polarity. The power supply voltage phase calculating unit  23  calculates the sinusoidal value sin θ{circumflex over ( )} s  on the basis of the calculated power supply voltage phase estimation value θ{circumflex over ( )} s . Note that, in a case of implementing resetting of the power supply voltage phase estimation value θ{circumflex over ( )} s  by using an interrupt function of a microcomputer, the power supply voltage phase calculating unit  23  resets the power supply voltage phase estimation value θ{circumflex over ( )} s  by using a signal output from a zero crossing detecting circuit as an interrupt signal. The zero crossing detecting circuit is a circuit that detects a timing at which the power supply voltage Vs switches from the negative polarity to the positive polarity. Note that the method for calculating the power supply voltage phase estimation value θ{circumflex over ( )} s  is not limited to the example described above, and any method may be used therefor. 
       FIG. 15  is a diagram illustrating an example of a configuration of the first pulse generating unit of the power converting apparatus according to the first embodiment. The first pulse generating unit  24  includes a carrier generating unit  241 , a reference PWM generating unit  242 , a dead time generating unit  243 , and a pulse selector  244 . 
     The carrier generating unit  241  generates a carrier wave carry, which is a carrier signal. The carrier wave carry is used for generation of a reference PWM signal Scom. An example of the carrier wave carry can be a triangular wave with a peak value “1” and a trough value “0”. The reference PWM signal Scom is a signal that is a reference of PWM signals used for driving the switching elements  311 ,  312 ,  321 , and  322 . As described above, in the first embodiment, complementary PWM control is assumed, in which a reference PWM signal is used for driving one of the switching elements of the first arm  31 , and a PWM signal complementary to the reference PWM signal is used for the other of the switching elements of the first arm  31 . 
     The reference PWM generating unit  242  compares the magnitudes of the reference ON-duty duty calculated by the ON-duty control unit  22  illustrated in  FIG. 13  and the carrier wave carry to generate the reference PWM signal Scom.  FIG. 16  is a chart illustrating an example of the reference ON-duty, the carrier wave, and the reference PWM signal in  FIG. 15 . As illustrated in  FIG. 16 , the reference PWM generating unit  242  generates the reference PWM signal Scom in such a manner that the reference PWM signal Scom has a value representing ON in the case where reference ON-duty duty&gt;carrier wave carry, and that the reference PWM signal Scom has a value representing OFF in the case where reference ON-duty duty&lt;carrier wave carry.  FIG. 16  illustrates a high active reference PWM signal Scom as an example. The high active reference PWM signal Scom is a signal with a high level representing ON and a low level representing OFF. Note that the signal generated by the reference PWM generating unit  242  is not limited to a high active reference PWM signal Scom, and may be a low active reference PWM signal Scom. The low active reference PWM signal Scom is a signal with a high level representing OFF and a low level representing ON. 
     The description refers back to  FIG. 15 , in which the dead time generating unit  243  generates a first PWM signal Sig 1  and a second PWM signal Sig 2 , which are two complementary signals, on the basis of the reference PWM signal Scom, and outputs the first PWM signal Sig 1  and the second PWM signal Sig 2 . Specifically, the dead time generating unit  243  generates an inverted PWM signal Scom′ that is a signal obtained by inverting the reference PWM signal Scom. The dead time generating unit  243  then generates the first PWM signal Sig 1  and the second PWM signal Sig 2  by setting a dead time in the reference PWM signal Scom and the inverted PWM signal Scom′. 
     Specifically, the dead time generating unit  243  generates the first PWM signal Sig 1  and the second PWM signal Sig 2  such that the first PWM signal Sig 1  and the second PWM signal Sig 2  both have a value representing OFF during the dead time. In one example, the dead time generating unit  243  makes the first PWM signal Sig 1  identical to the reference PWM signal Scom. In addition, the dead time generating unit  243  generates the second PWM signal Sig 2  by changing a signal value of the inverted PWM signal Scom′ from a value representing ON to a value representing OFF during the dead time. 
     In the case where the inverted PWM signal Scom′ is generated by inversion of the reference PWM signal Scom and two switching elements constituting one arm are respectively driven by the reference PWM signal Scom and the inverted PWM signal Scom′, there is, ideally, no period during which two switching elements constituting one arm are ON at the same time. Typically, however, a delay occurs in a transition from an ON state to an OFF state, and a delay occurs in a transition from an OFF state to an ON state. Thus, the delays result in a period during which two switching elements constituting one arm are ON at the same time, and may cause short-circuit of the two switching elements constituting one arm. The dead time is a period set such that two switching elements constituting one arm are not on at the same time even when a delay in a state transition occurs. During the dead time, two PWM signals for driving the two switching elements constituting one arm are both set to a value representing OFF. 
       FIG. 17  is a chart illustrating an example of the reference PWM signal, the inverted PWM signal, the first PWM signal, and the second PWM signal in  FIG. 15 .  FIG. 17  illustrates the reference PWM signal Scom, the inverted PWM signal Scom′, the first PWM signal Sig 1 , and the second PWM signal Sig 2  in this order from the top. In  FIG. 17 , when the inverted PWM signal Scom′ has a value representing ON, the second PWM signal Sig 2  has a value representing OFF during a dead time td. Note that the method for generating the dead time td described above is an example, the method for generating the dead time td is not limited to the above-described example, and any method may be used therefor. 
     The description refers back to  FIG. 15 , in which the pulse selector  244  determines which of the driving circuits for the switching element  311  and the switching element  312  to transmit each of the first PWM signal Sig 1  and the second PWM signal Sig 2  output from the dead time generating unit  243 .  FIG. 18  is a flowchart illustrating an example of procedures of a selecting process performed by the pulse selector of the first pulse generating unit illustrated in  FIG. 15 . The pulse selector  244  first determines whether or not the polarity of the power supply voltage Vs is positive, that is Vs&gt;0 (step S 1 ). If the polarity of the power supply voltage Vs is positive (step S 1 : Yes), the pulse selector  244  transmits the first PWM signal Sig 1  as pulse_ 312 A to the driving circuit for the switching element  312 , and transmits the second PWM signal Sig 2  as pulse_ 311 A to the driving circuit for the switching element  311  (step S 2 ). This is because, when the power supply voltage Vs is positive, the current path is switched between the current path illustrated in  FIG. 5  and the current path illustrated in  FIG. 3  by turning OFF or ON of the switching element  311  and the switching element  312 , that is, the bus voltage Vdc and the power supply current Is are controlled by switching operation of the switching element  311  and the switching element  312 . 
     If the polarity of the power supply voltage Vs is negative (step S 1 : No), the pulse selector  244  transmits the first PWM signal Sig 1  as pulse_ 311 A to the driving circuit for the switching element  311 , and transmits the second PWM signal Sig 2  as pulse_ 312 A to the driving circuit for the switching element  312  (step S 3 ). This is because, when the power supply voltage Vs is negative, the current path is switched between the current path illustrated in  FIG. 6  and the current path illustrated in  FIG. 4  by turning OFF or ON of the switching element  311  and the switching element  312 , that is, the bus voltage Vdc and the power supply current Is are controlled by switching operation of the switching element  311  and the switching element  312 . The pulse selector  244  repeats the above-described operation each time the polarity of the power supply voltage Vs changes. 
     As described above, the first pulse generating unit  24  generates pulse_ 311 A that is a signal for driving the switching element  311  and pulse_ 312 A that is a signal for driving the switching element  312 . 
     As described above, because the switching element  311  and the switching element  312  are complementarily controlled, the process of generating the inverted PWM signal Scom′ from the reference PWM signal Scom can be achieved by using a simple signal inversion process. In addition, the relation of driving pulse outputs in one carrier can be made approximately the same regardless of the power supply voltage polarity, and prevention of a short circuit of the upper and lower arms can be easily achieved. Stable control can be achieved by simple processes. 
     In addition, in the power converting apparatus  100  according to the first embodiment, synchronous rectification control by the switching elements  311  and  312  of the first arm  31  can be achieved. Thus, in the power converting apparatus  100  according to the first embodiment, loss can be reduced in a region in which the loss of a switching element is smaller than the loss of a body diode, that is, a region in which each of currents flowing through the switching element and the body diode is small as illustrated in  FIG. 19 . Thus, a highly-efficient system can be achieved. 
       FIG. 19  is a schematic graph illustrating the relation of currents flowing through a switching element and a body diode illustrated in  FIG. 1 , the loss of the switching element, and the loss of the body diode. The horizontal axis in  FIG. 19  represents a current flowing through the switching element in the ON state, and a current flowing through the body diode. The vertical axis in  FIG. 19  represents a loss caused when the current flows through the switching element in the ON state and a loss caused when the current flows through the body diode. A solid line depicts the loss characteristics of the body diode. The loss characteristics of the body diode indicate the relation between the current flowing through the body diode and the loss caused by the ON-resistance of the body diode when the current flows. A dotted line depicts the loss characteristics of the switching element in the ON state. The loss characteristics indicate the relation between the current flowing through the carrier of the switching element and the loss caused by the ON-resistance of the switching element when the current flows. A region represented by a sign A is a region in which the currents flowing through the switching element and the body diode are small. A region represented by a sign B is a region in which the currents flowing through the switching element and the body diode are large. At the boundary between the region A and the region B, the currents are equal to a current value at which the value of the loss caused in the switching element and the value of the loss caused in the body diode are equal. 
     As illustrated in  FIG. 19 , in the region B in which the loss of the switching element is higher than the loss of the body diode, the complementary operation is stopped, so that an increase in the loss due to the synchronous rectification control can be prevented or reduced. Thus, by the control of switching between performing and not performing the synchronous rectification control depending on the power supply current Is, a highly efficient system can be achieved in all load regions. 
     Note that optimal values according to a driving condition are present for control parameters used for computation by the power supply current command value control unit  21  and the ON-duty control unit  22  illustrated in  FIG. 13 . The driving condition is expressed by at least one value of the power supply voltage Vs, the power supply current Is, and the bus voltage Vdc. For example, it is desirable that a proportional control gain in the ON-duty control unit  22  change in inverse proportion to the bus voltage Vdc. This is because, if the value of a control parameter is constant with respect to a change in in the driving condition, the control parameter will significantly deviate from a value suitable for control, and as a result, harmonics of the power supply current Is may increase, pulsation of the bus voltage Vdc may increase, and power-supply power factor may decrease. In order to prevent or reduce such increase in pulsation of the bus voltage Vdc, decrease in the power-supply power factor, and the like, the power supply current command value control unit  21  and the ON-duty control unit  22  may hold a calculation formula or a table for implementing a desired circuit operation, and adjust a control parameter on the basis of detected information by using the calculation formula or the table. The configuration to adjust a control parameter on the basis of detected information makes the control parameter a value suitable for control, which improves controllability. Note that the detected information is at least one of the power supply voltage Vs, the power supply current Is, and the bus voltage Vdc, or information from which these values can be estimated, for example. An example of the information from which the values can be estimated is power information detected by a detector for detecting a power supplied from the alternating-current power supply  1 . 
     In addition, while the proportional-integral control is presented as the computation method used in the power supply current command value control unit  21  and the ON-duty control unit  22  in the example described above, the present invention is not limited to this computation method, and other computation methods may be used and a derivative term may be added to perform proportional-integral-derivative control. In addition, the computation methods in the power supply current command value control unit  21  and the ON-duty control unit  22  need not be the same computation method. 
     The description refers back to  FIG. 13 , in which the second pulse generating unit  25  generates pulse_ 321 A that is a signal for driving the switching element  321  and pulse_ 322 A that is a signal for driving the switching element  322  on the basis of the power supply voltage Vs detected by the power supply voltage detecting unit  5  and the power supply current Is detected by the power supply current detecting unit  6 , and outputs the pulse_ 321 A and the pulse_ 322 A. 
       FIG. 20  is a flowchart illustrating an example of procedures of a process performed by the second pulse generating unit illustrated in  FIG. 13 . A basic operation of the second pulse generating unit  25  is controlling the ON or OFF states of the switching element  321  and the switching element  322  depending on the polarity of the power supply voltage Vs. As illustrated in  FIG. 20 , the second pulse generating unit  25  determines whether or not the polarity of the power supply voltage Vs is positive, that is, Vs&gt;0 (step S 11 ). If the polarity of the power supply voltage Vs is positive (step S 11 : Yes), the second pulse generating unit  25  generates and outputs pulse_ 321 A and pulse_ 322 A to turn the switching element  321  OFF and turn the switching element  322  ON (step S 12 ). 
     If the polarity of the power supply voltage Vs is negative (step S 11 : No), the second pulse generating unit  25  generates and outputs pulse_ 321 A and pulse_ 322 A to turn the switching element  321  ON and turn the switching element  322  OFF (step S 13 ). This enables the synchronous rectification control, and a highly efficient system can be achieved as described above. 
     As described above, however, when the switching element  311  and the switching element  322  are turned ON while the power supply current Is is not flowing, a capacitor short circuit via the alternating-current power supply  1  and the reactor  2  occurs. Thus, in addition to control of the switching element  311  and the switching element  322 , the power converting apparatus  100  according to the first embodiment controls the ON or OFF states of the switching element  321  and the switching element  322  on the basis of the power supply current Is. 
       FIG. 21  is a flowchart illustrating an example of procedures for controlling the switching elements on the basis of the power supply current by the second pulse generating unit illustrated in  FIG. 13 . As illustrated in  FIG. 21 , it is determined whether or not the absolute value of the power supply current Is is larger than the current threshold β (step S 21 ). If the absolute value of the power supply current Is is larger than the current threshold β (step S 21 : Yes), the second pulse generating unit  25  permits the switching element  321  and the switching element  322  to be ON (step S 22 ). When the switching element  321  and the switching element  322  are permitted to be ON, the ON and OFF states are controlled depending on the polarity of the power supply voltage Vs illustrated in  FIG. 20 . 
     If the absolute value of the power supply current Is is equal to or smaller than the current threshold β (step S 21 : No), the second pulse generating unit  25  does not permit the switching element  321  and the switching element  322  to be ON (step S 23 ). When the switching element  321  and the switching element  322  are not permitted to be ON, the switching element  321  and the switching element  322  are controlled to be in the OFF state regardless of the polarity of the power supply voltage Vs illustrated in  FIG. 20 . 
     As a result of the control described above, when a current larger than the current threshold Q flows in the forward direction in the body diodes of the switching elements, the switching element  321  and the switching element  322  are turned ON. This enables prevention of a capacitor short circuit via the alternating-current power supply  1  and the reactor  2 . In addition, the second pulse generating unit  25  may control the switching element  321  and the switching element  322  by using the polarity of the power supply current Is, that is, the direction in which the current flows, instead of ON-OFF control depending on the polarity of the power supply voltage Vs. 
     In addition, instead of the process illustrated in  FIG. 21 , whether or not to permit the switching element  321  and the switching element  322  to be ON may be determined on the basis of the state of switching control. When switching is not performed, no current flows in the switching elements, and thus a timing to enter such a state is predicted so as not to permit the switching element  321  and the switching element  322  to be ON. Note that, in this case, the synchronous rectification effect may not be produced in a state in which passive full-wave rectification, that is, a short-circuit path is not used, but control can be simply built independently of detection of a current or a voltage. 
     In addition, whether or not to permit the switching element  321  and the switching element  322  to be ON may be determined on the basis of a difference between the power supply voltage Vs and the bus voltage Vdc instead of the process illustrated in  FIG. 21 . Specifically, if (power supply voltage−bus voltage)&gt;0, the switching element  321  and the switching element  322  are permitted to be ON, and if (power supply voltage−bus voltage)&gt;0, the switching element  321  and the switching element  322  are not permitted to be ON. 
     Note that, in the example described above, the second pulse generating unit  25  selects the switching element to be turned ON from the switching element  321  and the switching element  322  on the basis of the power supply voltage polarity, and controls switching element  321  and the switching element  322  on the basis of the power supply current Is to prevent a capacitor short circuit. The control, however, is not limited to this example, and the first pulse generating unit  24  may determine whether or not to permit the switching elements  311 ,  312 ,  321 , and  322  to be ON on the basis of the power supply current Is to prevent a capacitor short circuit, and the second pulse generating unit  25  may perform switching depending on the power supply voltage polarity without performing control to prevent a capacitor short circuit on the switching element  321  and the switching element  322 . 
     Specifically, in the case where the power supply voltage Vs is positive, the first pulse generating unit  24  does not permit the switching element  311  to be ON when the absolute value of the power supply current Is is equal to or smaller than the current threshold Q, and permits the switching element  311  to be ON when the absolute value of the power supply current Is is larger than the current threshold β. In contrast, in the case where the power supply voltage Vs is negative, the first pulse generating unit  24  does not permit the switching element  312  to be ON when the absolute value of the power supply current Is is equal to or smaller than the current threshold @, and permits the switching element  312  to be ON when the absolute value of the power supply current Is is larger than the current threshold β. 
     In addition, while the switching in each of the arms in each power supply cycle is achieved by the method of generating complementary PWM signals in the example described above, the method of generating PWM signals is not limited to this example. Specifically, the control unit  10  may generate a signal pulse_ 312 A for driving the switching element  312  when the power supply voltage Vs is positive, and generate a signal pulse_ 311 A for driving the switching element  311  when the power supply voltage Vs is negative. In addition, in this case, the control unit  10  may generate PWM signals for driving the switching elements  311  and  312  on the basis of the relation of the power supply current Is, the power supply voltage Vs, and the bus voltage Vdc. This enables the switching elements  311  and  312  to be turned OFF before the timing at which the power supply current Is becomes zero, and in this case, a capacitor short circuit via the alternating-current power supply  1  and the reactor  2  can be prevented even when the operations of the switching elements  321  and  322  are controlled on the basis of the power supply voltage polarity. 
       FIG. 22  is a chart illustrating a first example of signals, corresponding to one cycle of the power supply voltage, generated in the power converting apparatus according to the first embodiment.  FIG. 22  illustrates an example of the signals generated by the process explained with reference to  FIG. 20 . In  FIG. 22 , the horizontal axis represents time, and the power supply voltage Vs, the power supply current Is, a timer set value α and a carrier signal, a signal for driving the switching element  311 , a signal for driving the switching element  312 , a signal for driving the switching element  321 , and a signal for driving the switching element  322  are illustrated in this order from the top. 
     The timer set value a is a command value associated with the reference ON-duty duty, and changes with time in a stepwise manner. The timer set value α is a period with the value of each step in the vertical axis being constant. The reference ON-duty duty associated with each timer set value a changing in such a stepwise manner is compared with the carrier wave carry that is the carrier signal, and the pulse widths of the switching elements  311  and  321  are thus determined. The reference ON-duty duty is small near the zero crossing of the power supply voltage Vs, and becomes larger as the power supply voltage Vs approaches its peak value. Note that the dead time is not illustrated in  FIG. 22 . 
     A current threshold (positive) on the positive side is set to prevent excessive switching operations near the zero crossing when the power supply current Is changes from negative to positive. Similarly, a current threshold (negative) on the negative side is set to prevent excessive switching operations near the zero crossing when the power supply current Is changes from positive to negative. 
       FIG. 22  illustrates an example of operations for complementarily performing PWM control on the switching elements  311  and  312 , in which the switching element  312  is a master when the power supply voltage Vs has the positive polarity and the switching element  311  is a master when the power supply voltage Vs has the negative polarity. Thus, a reference ON-duty duty of an arc shape that is convex downward is used when the power supply voltage Vs has the positive polarity, and a reference ON-duty duty of an arc shape that is convex downward is also used when the power supply voltage Vs has the negative polarity. 
     The switching elements  321  and  322  are switched ON or OFF depending on the polarity of the power supply voltage Vs, and are further turned OFF when the absolute value of the power supply current Is is equal to or smaller than the current threshold. Note that the power converting apparatus  100  according to the first embodiment may have a configuration in which the power supply current detecting unit  6  has a filter or hysteresis to prevent excessive switching operations near the current thresholds. Alternatively, the power converting apparatus  100  according to the first embodiment may have a configuration in which the control unit  10  has a filter to the power supply current Is or hysteresis to prevent excessive switching operations near the current thresholds. 
       FIG. 23  is a chart illustrating a second example of signals, corresponding to one cycle of the power supply voltage, generated in the power converting apparatus according to the first embodiment. In  FIG. 23 , in a manner similar to  FIG. 22 , the horizontal axis represents time, and the power supply voltage Vs, the power supply current Is, the timer set value α and the carrier signal, a signal for driving the switching element  311 , a signal for driving the switching element  312 , a signal for driving the switching element  321 , and a signal for driving the switching element  322  are illustrated in this order from the top. 
       FIG. 23  illustrates an example of operations for complementarily performing PWM control on the switching elements  311  and  312 , in which the switching element  312  is a master when the power supply voltage Vs has the positive polarity and when the power supply voltage Vs has the negative polarity. Thus, a reference ON-duty duty of an arc shape that is convex downward is used when the power supply voltage Vs has the positive polarity, and a reference ON-duty duty of an arc shape that is convex upward is used when the power supply voltage Vs has the negative polarity. In the example of the operations in  FIG. 23 , the signal pulse_ 312 A for driving the switching element  312  is generated when the power supply voltage Vs has the positive polarity, and the signal pulse_ 311 A for driving the switching element  311  is generated when the power supply voltage Vs has the negative polarity. 
     In addition, while the example in which the switching elements are controlled by the carrier signals is presented in  FIG. 22  described above, the operations of the first embodiment are also applicable to simple switching control in which switching is performed once to several times during a half cycle of the power supply cycle.  FIG. 24  is a chart illustrating an example of signals when the power converting apparatus according to the first embodiment performs simple switching control. In  FIG. 24 , the horizontal axis represents time, and the power supply voltage Vs, the power supply current Is, the absolute value Is of the power supply current Is, a power supply polarity signal, a power supply current signal, a signal for driving the switching element  311 , a signal for driving the switching element  312 , a signal for driving the switching element  321 , and a signal for driving the switching element  322  are illustrated in this order from the top. The power supply polarity signal is a binary signal that changes with the polarity of the power supply voltage Vs, and is used for controlling the switching element operations of the switching elements  311  and  312 . The power supply current signal is a binary signal used for controlling the switching element operations of the switching elements  321  and  322 . 
     In  FIG. 24 , three current thresholds are illustrated. A current threshold on the positive side of the power supply current Is is a threshold set for a purpose similar to that of the current threshold (positive) on the positive side described with reference to  FIG. 22 . A current threshold on the negative side of the power supply current Is is a threshold set for a purpose similar to that of the current threshold (negative) on the negative side described with reference to  FIG. 22 . A current threshold set for the absolute value Isi of the power supply current Is is a threshold set for changing the value of the power supply current signal. 
     The power supply polarity signal is generated by detection of the zero crossing of the power supply voltage Vs, and the power supply current signal is generated by detection of the zero crossing of the power supply current Is. In this case, when the absolute value |Is| of the power supply current Is is equal to or smaller than the current threshold, the power converting apparatus  100  performs control such that the switching element  311  and the switching element  321  are not ON at the same time and such that the switching element  312  and the switching element  322  are not ON at the same time. This enables prevention of a capacitor short circuit. 
     In addition, even when the switching elements  311  and  312  are in a passive state in which no switching operations are performed, the switching element  321  and the switching element  322  are prevented from being turned ON when the absolute value of the power supply current Is is equal to or smaller than the current threshold, which enables prevention of a capacitor short circuit. 
       FIG. 25  is a chart illustrating an example of signals in passive states generated by the power converting apparatus according to the first embodiment. In  FIG. 25 , in a manner similar to  FIG. 24 , the horizontal axis represents time, and the power supply voltage Vs, the power supply current Is, the absolute value Isi of the power supply current Is, a power supply polarity signal, a power supply current signal, a signal for driving the switching element  311 , a signal for driving the switching element  312 , a signal for driving the switching element  321 , and a signal for driving the switching element  322  are illustrated in this order from the top. In this case as well, when the absolute value of the power supply current Is equal to or smaller than the current threshold, the power converting apparatus  100  performs control such that the switching element  311  and the switching element  321  are not ON at the same time and such that the switching element  312  and the switching element  322  are not ON at the same time. This enables prevention of a capacitor short circuit. 
     Next, driving circuits and bootstrap circuits for the switching elements will be described with reference to  FIGS. 26 to 29 . 
       FIG. 26  is a diagram illustrating driving circuits and bootstrap circuits included in the power converting apparatus according to the first embodiment. As illustrated in  FIG. 26 , the power converting apparatus  100  includes two direct-current voltage sources  300 , four driving circuits  311 DC,  312 DC,  321 DC, and  322 DC, and two bootstrap circuits  401  and  402 , in addition to the configuration illustrated in  FIG. 1 . While the driving circuits  311 DC and  312 DC share one direct-current voltage source  300  and the driving circuits  321 DC and  322 DC share the other direct-current voltage source  300  in the power converting apparatus  100  of  FIG. 26 , one direct-current voltage source  300  may be used instead of the two direct-current voltage sources  300 , and the four driving circuits  311 DC,  312 DC,  321 DC, and  322 DC may share the one direct-current voltage source  300 . 
     The driving circuit  311 DC that is a first driving circuit converts pulse_ 311 A from the control unit  10  into a first driving signal for driving the switching element  311  by using a voltage output from the bootstrap circuit  401  as the power supply voltage, and outputs the first driving signal to the gate of the switching element  311 . Details of the configuration of the bootstrap circuit  401  will be described later. The driving circuit  312 DC that is a second driving circuit converts pulse_ 312 A from the control unit  10  into a second driving signal for driving the switching element  312  by using a voltage output from the direct-current voltage source  300  as the power supply voltage, and outputs the second driving signal to the gate of the switching element  312 . 
     The driving circuit  321 DC converts pulse_ 321 A from the control unit  10  into a driving signal for driving the switching element  321  by using a voltage from the bootstrap circuit  402  as the power supply voltage, and outputs the driving signal to the gate of the switching element  321 . The driving circuit  322 DC converts pulse_ 322 A from the control unit  10  into a driving signal for driving the switching element  322  by using a voltage output from the direct-current voltage source  300  as the power supply voltage, and outputs the driving signal to the gate of the switching element  322 . 
     The bootstrap circuit  401  includes a boot resistor  311 R having one end connected to the direct-current voltage source  300 , a boot diode  311 D having an anode connected to the other end of the boot resistor  311 R, a boot capacitor  311 C that is a second capacitor having one end connected to a cathode of the boot diode  311 D and the other end connected to the driving circuit  311 DC, and a gate voltage suppression diode  311 D′. 
     An anode of the gate voltage suppression diode  311 D′ is connected to the cathode of the boot diode  311 D and one end of the boot capacitor  311 C. A cathode of the gate voltage suppression diode  311 D′ is connected to the driving circuit  311 DC. Assume that the value of a first voltage that is a voltage at which a forward current starts to flow in the gate voltage suppression diode  311 D′ is lower than the value of a second voltage that is a voltage at which a forward current starts to flow in the body diode  312   a . Thus, assume that the forward current-forward voltage characteristics of the gate voltage suppression diode  311 D′ are superior to the forward current-forward voltage characteristics of the body diode  312   a . Note that, a voltage at which a forward current starts to flow in a diode is typically called a forward voltage. 
     The bootstrap circuit  402  has a configuration similar to that of the bootstrap circuit  401 , and includes a boot resistor  321 R having one end connected to the direct-current voltage source  300 , a boot diode  321 D having an anode connected to the other end of the boot resistor  321 R, a boot capacitor  321 C having one end connected to a cathode of the boot diode  321 D and the other end connected to the driving circuit  321 DC, and a gate voltage suppression diode  321 D′. 
     An anode of the gate voltage suppression diode  321 D′ is connected to the cathode of the boot diode  321 D and one end of the boot capacitor  321 C. A cathode of the gate voltage suppression diode  321 D′ is connected to the driving circuit  321 DC. Assume that the value of a voltage at which a forward current starts to flow in the gate voltage suppression diode  321 D′ is lower than the value of a voltage at which a forward current starts to flow in the body diode  322   a . Thus, assume that the forward current-forward voltage characteristics of the gate voltage suppression diode  321 D′ are superior to the forward current-forward voltage characteristics of the body diode  322   a . The reason why the gate voltage suppression diode  311 D′ is used will be described later. Note that, because the bootstrap circuit  402  has a configuration similar to that of the bootstrap circuit  401 , details of the configuration of the bootstrap circuit  402  will not be described. 
     In the bootstrap circuit  401  having such a configuration, when the switching element  312  is turned ON, a current flows through a path constituted by the direct-current voltage source  300 , the boot resistor  311 R, the boot diode  311 D, the boot capacitor  311 C, and the switching element  312 , and the boot capacitor  311 C is charged. A capacitor voltage V c  generated across the ends of the charged boot capacitor  311 C can be expressed as V c   m =V dc +V BD −V dr −V f . V dc  represents the voltage of the direct-current voltage sources  300 , V BD  represents the forward voltage of the body diode  312   a , V dr  represents a drop voltage of the boot resistor  311 R, and V f  represents the forward voltage of the boot diode  311 D. 
     For example, when V dc  is 6.0 V, V BD  is 3.0 V, V dr  is 0.5 V, and V f  is 1.5 V, V c  is 7.0 V. In this case, when the rated voltage of the driving circuit  311 DC is 6.0 V, the value of V c  is higher than the rated voltage of the driving circuit  311 DC. The reason why the value of V c  is high is that the forward voltage of the body diode  312   a  is also applied to the boot capacitor  311 C in addition to the voltage of the direct-current voltage source  300 . The forward voltage of the body diode  312   a  is a voltage at which a forward current starts to flow in the body diode  312   a . For example, in a case where a switching element made of a WBG semiconductor in which the potential barrier of a p-n junction is high is used as the switching element  312 , the forward voltage of the body diode  312   a  of the switching element  312  tends to be high. Note that the switching element  312  in which the forward voltage of the body diode  312   a  becomes high is not limited to a switching element made of a WBG semiconductor, and a Si switching element in which the forward voltage of a body diode tends to be high so that the capacitor voltage V c  of the boot capacitor  311 C is higher than the rated voltage of the driving circuit  311 DC may also be applicable. 
     When the capacitor voltage V c  becomes higher than the rated voltage of the driving circuit  311 DC, the withstand voltage of the driving circuit  311 DC may decrease. In addition, because the value of the driving signal generated by the driving circuit  311 DC becomes larger, the short circuit withstand of the switching element  311  may decrease. In addition, when the switching element  311  is driven by the driving circuit  311 DC to which such a high voltage is applied, the value of the driving signal generated by the driving circuit  311 DC becomes larger than the value of the driving signal generated by the driving circuit  312 DC to which the voltage of the direct-current voltage source  300  is applied. Thus, the value of the loss of the switching element  311  in the ON state and the value of the loss of the switching element  312  in the ON state are different from each other, and imbalance in heat generation between the switching element  311  and the switching element  312  increases. When the imbalance in heat generation increases and a junction temperature of a semiconductor constituting one of the switching elements exceeds a permissible value, there is a possibility that normal operations can no longer be performed. 
     In the power converting apparatus  100  illustrated in  FIG. 26 , the gate voltage suppression diode  311 D′ is provided between the boot capacitor  311 C and the driving circuit  311 DC. In other words, the boot capacitor  311 C is connected with the driving circuit  311 DC via the gate voltage suppression diode  311 D′. Thus, the capacitor voltage of the boot capacitor  311 C is reduced by a certain value by the gate voltage suppression diode  311 D′, and then applied as the power supply voltage for the driving circuit  311 DC to the driving circuit  311 DC. In this manner, the gate voltage suppression diode  311 D′ functions as a power supply voltage adjusting element for adjusting the power supply voltage for the driving circuit  311 DC to be applied from the boot capacitor  311 C to the driving circuit  311 DC. 
     For example, when V dc  is 6.0 V, V BD  is 3.0 V, V dr  is 0.5 V, V f  is 1.5 V, and V D  is 1.0 V, the value of V c  is expressed as V c =V dc +V BD −V dr −V f −V D  and V c =6.0 V is obtained. V D  represents a forward voltage of the gate voltage suppression diode  311 D′, that is, a voltage at which a forward current starts to flow in the gate voltage suppression diode  311 D′. 
     As described above, while a driving voltage equal to V c  (7.0 V) is applied to the driving circuit  311 DC when the gate voltage suppression diode  311 D′ is not provided, 6.0 V is applied to the driving circuit  311 DC when the gate voltage suppression diode  311 D′ is provided. Thus, as a result of providing the gate voltage suppression diode  311 D′, the power supply voltage for the driving circuit  311 DC to be applied from the boot capacitor  311 C to the driving circuit  311 DC can be reduced to the rated voltage of the driving circuit  311 DC. In addition, when V dc , V BD , V dr , V f , V D , etc. are set as described above, the power supply voltage for the driving circuit  311 DC becomes equal to the voltage V dc  of the direct-current voltage source  300 . 
     According to the power converting apparatus  100  according to the first embodiment, a decrease in the withstand voltage of the driving circuit  311 DC can be prevented or reduced, and a decrease in the short circuit withstand of the switching element  311  can be prevented or reduced. In addition, because the power supply voltage for the driving circuit  311 DC can be adjusted to a value equal to the power supply voltage for the driving circuit  312 DC, the imbalance in heat generation between the switching element  311  and the switching element  312  can be reduced, which improves the reliability of the power converting apparatus  100 . 
     In addition, according to the power converting apparatus  100  according to the first embodiment, because the power supply voltage for the driving circuit  311 DC can be adjusted to a value equal to the power supply voltage for the driving circuit  312 DC, the driving circuit  311 DC and the driving circuit  312 DC can be constituted by common components. This improves the yield of components as compared with a case where the driving circuit  311 DC and the driving circuit  312 DC are produced from different components. In addition, the manufacturing cost of the driving circuit  311 DC and the driving circuit  312 DC is reduced, and the volume of the components during manufacture of the driving circuit  311 DC and the driving circuit  312 DC can be reduced. Furthermore, replacement of the driving circuit  311 DC and the driving circuit  312 DC in repairing the power converting apparatus  100  is facilitated. 
     Note that, while the gate voltage suppression diode  311 D′ is provided inside the bootstrap circuit  401  in the power converting apparatus  100  illustrated in  FIG. 26 , the gate voltage suppression diode  311 D′ may be produced separately from the bootstrap circuit  401  and provided between the bootstrap circuit  401  and the driving circuit  311 DC. In the case where the gate voltage suppression diode  311 D′ is provided inside the bootstrap circuit  401 , the bootstrap circuit  401  can be manufactured such that the gate voltage suppression diode  311 D′, the boot capacitor  311 C, and the like are formed integrally. This improves the production efficiency of the power converting apparatus  100 . In the case where the gate voltage suppression diode  311 D′ is produced separately from the bootstrap circuit  401  and provided between the bootstrap circuit  401  and the driving circuit  311 DC, a suitable gate voltage suppression diode  311 D′ according to the value of the forward voltage of the body diode  312   a  can be selected from a plurality of gate voltage suppression diodes  311 D′ with different forward voltages and can be mounted. This enables the power supply voltage for the driving circuit  311 DC to be easily adjusted. 
       FIG. 27  is a diagram illustrating an example of a configuration of a power converting apparatus according to a first modification of the first embodiment. In a power converting apparatus  100 - 1  illustrated in  FIG. 27 , bootstrap circuits  401 A and  402 A are used instead of the bootstrap circuits  401  and  402  illustrated in  FIG. 26 . In the bootstrap circuit  401 A, the gate voltage suppression diode  311 D′ is not provided, and one end of the boot capacitor  311 C is directly connected to the driving circuit  311 DC. In the bootstrap circuit  402 A, the gate voltage suppression diode  321 D′ is not provided, and one end of the boot capacitor  321 C is directly connected to the driving circuit  321 DC. In addition, in the power converting apparatus  100 - 1 , a gate voltage suppression diode  312 RD is connected in parallel with the switching element  312 , and a gate voltage suppression diode  322 RD is connected in parallel with the switching element  322 . 
     An anode of the gate voltage suppression diode  312 RD is connected to an anode of the body diode  312   a , and a cathode of the gate voltage suppression diode  312 RD is connected to a cathode of the body diode  312   a . Assume that the forward current-forward voltage characteristics of the gate voltage suppression diode  312 RD are superior to the forward current-forward voltage characteristics of the body diode  312   a . For example, when the forward voltage of the gate voltage suppression diode  312 RD is 1.5 V and the forward voltage of the body diode  312   a  is 3.0 V, the boot capacitor  311 C is charged by a voltage having a value obtained by subtracting the drop voltage of the boot resistor  311 R and the forward voltage of the boot diode  311 D from a sum of 1.5 V and the voltage of the direct-current voltage source  300 . The capacitor voltage of the charged boot capacitor  311 C has a value smaller than that in the case where no gate voltage suppression diode  312 RD is used, and is used as the power supply voltage for the driving circuit  311 DC. As described above, the gate voltage suppression diode  312 RD functions as a capacitor voltage adjusting element for adjusting a capacitor voltage generated across the ends of the boot capacitor  311 C. 
     An anode of the gate voltage suppression diode  322 RD is connected to an anode of the body diode  322   a , and a cathode of the gate voltage suppression diode  322 RD is connected to a cathode of the body diode  322   a . Assume that the forward current-forward voltage characteristics of the gate voltage suppression diode  322 RD are superior to the forward current-forward voltage characteristics of the body diode  322   a . The gate voltage suppression diode  322 RD functions as a capacitor voltage adjusting element for adjusting the capacitor voltage generated across the ends of the boot capacitor  321 C. 
     According to the power converting apparatus  100 - 1  illustrated in  FIG. 27 , an increase in the charging voltage of the boot capacitor can be prevented or reduced, and an increase in the loss due to the body diode during an asynchronous rectification period of the zero crossing and the dead time can be prevented or reduced. 
       FIG. 28  is a diagram illustrating an example of a configuration of a power converting apparatus according to a second modification of the first embodiment. In a power converting apparatus  100 - 2  illustrated in  FIG. 28 , the bootstrap circuit  402 A illustrated in  FIG. 27  is used instead of the bootstrap circuit  402  illustrated in  FIG. 26 . Thus, in the power converting apparatus  100 - 2 , the gate voltage suppression diode  311 D′ is used only in the first arm. 
     In a power converting apparatus having a full-bridge configuration like the power converting apparatus  100 - 2 , no path for charging the boot capacitor via the body diode occurs by synchronous rectification control based on the power supply polarities of the switching elements  321  and  322 . Thus, in the power converting apparatus  100 - 2 , the gate voltage suppression diode  311 D′ may be implemented only in the first arm, which enables reduction in used components. 
       FIG. 29  is a diagram illustrating an example of a configuration of a power converting apparatus according to a third modification of the first embodiment. In a power converting apparatus  100 - 3  illustrated in  FIG. 29 , the gate voltage suppression diode  322 RD illustrated in  FIG. 27  is not provided. Thus, in the power converting apparatus  100 - 3 , the gate voltage suppression diode  312 RD is used only in the first arm. In the power converting apparatus  100 - 3 , in a manner similar to the power converting apparatus  100 - 2 , no path for charging the boot capacitor via the body diode occurs by synchronous rectification control based on the power supply polarities of the switching elements  321  and  322 . Thus, in the power converting apparatus  100 - 3 , the gate voltage suppression diode  312 RD may be implemented only in the first arm, which enables reduction in used components. 
     Note that, in the first embodiment, in a case where the alternating-current power supply  1  is a commercial power supply of 50 Hz or 60 Hz, the audible frequency is in a range from 16 kHz to 20 kHz, that is, a range from 266 to 400 times the frequency of the commercial power supply. When the switching elements are driven with such audible frequency, there is a problem of noise caused by switching. Because switching elements made of WBG semiconductors can perform fast switching, switching elements made of WBG semiconductors are suitable for switching elements that can be switched at a frequency higher than such audible frequency, such as a switching frequency higher than 20 kHz. 
     In addition, in a case where switching elements made of Si semiconductors are driven at a switching frequency of several tens of kHz or higher, such as a switching frequency higher than 20 kHz, the ratio of the switching loss increases, and a measure for heat radiation is essential. In the case of switching elements made of WBG semiconductors, the switching loss is much smaller than that in the case of switching elements made of Si semiconductors even when the switching elements are driven at a switching frequency higher than 20 kHz. Thus, use of switching elements made of WBG semiconductors in the power converting apparatus  100  eliminates the need for a measure for heat radiation of switching elements or allows miniaturization of members, such as radiating fins, used for a measure for heat radiation of switching elements, which enables reduction in size and weight of the power converting apparatus  100 . In addition, high-frequency switching of switching elements made of WBG semiconductors can be performed, which can make the inductance of the reactor  2  relatively smaller. Thus, the reactor  2  can be reduced in size. Note that the switching frequency is preferably equal to or lower than 150 kHz so that the primary component of the switching frequency is not included in a range of measurement of noise terminal voltage standard. 
     In addition, WBG semiconductors have a smaller capacitance than Si semiconductors; therefore, a recovery current caused by switching is low and the occurrence of a loss and noise caused by a recovery current can thus be reduced. Thus, WBG semiconductors are suitable for high-frequency switching. 
     In addition, even in a case where WBG semiconductors are driven at a high frequency about 100 kHz, an increase in a loss generated in the switching elements is prevented or reduced; therefore, the loss reduction effect produced by miniaturization of the reactor  2  increases. Thus, a highly efficient converter can be achieved in a wide output band, that is, under a wide load condition. 
     In addition, WBG semiconductors have a higher heat resistance than Si semiconductors, and have a higher permissible level of heat generation by switching due to imbalance in the loss between arms. Because the first arm  31  is driven at a higher frequency than the second arm  32  and the switching loss and the heat generation of the first arm  31  thus increase, WBG semiconductors are more suitable for the first arm  31  with high heat generation than the second arm  32 . 
     Note that super junction (SJ)-MOSFETs may be used for switching elements constituting an arm that performs slow switching. Use of SJ-MOSFETs for an arm that performs slow switching can reduce the disadvantages of SJ-MOSFETs, which are high capacitance and high occurrence of recovery, while making use of low ON-resistance that is an advantage of SJ-MOSFETs. In addition, use of SJ-MOSFETs can reduce the manufacturing cost of the arm that performs slow switching as compared to use of switching elements made of WBG semiconductors. 
     Note that the power converting apparatus  100  according to the first embodiment may be constituted by a general-purpose intelligent power module (IPM). Use of an IPM enables the driving circuits for the switching elements  311 ,  312 ,  321 , and  322  to be contained inside the IPM, which can reduce the board area on which the reactor  2 , the bridge circuit  3 , the smoothing capacitor  4 , the power supply voltage detecting unit  5 , the power supply current detecting unit  6 , the bus voltage detecting unit  7 , and the control unit  10  are mounted. In addition, use of a general-purpose IPM can prevent or reduce an increase in cost. 
     Note that the power converting apparatus  100  according to the first embodiment only needs to obtain the polarity of the power supply voltage Vs, and is not limited to the configuration for determining the polarity of the power supply voltage Vs by detecting a zero crossing point of the power supply voltage Vs. In the case of detecting a zero crossing point, in order to prevent erroneous determination of the polarity near the zero crossing, the power converting apparatus  100  turns the operations of the first arm  31  and the second arm  32  OFF for a predetermined period from the zero crossing point on the basis of the power supply voltage phase estimation value θ{circumflex over ( )} s . 
     While the switching element  321  and the switching element  322  are permitted to be in the ON state when the absolute value of the power supply current Is is equal to or larger than the current threshold in the power converting apparatus  100  according to the first embodiment, the configuration of the power converting apparatus  100  is not limited thereto. The power converting apparatus  100  may estimate that a current flows in a body diode of a switching element by using any of the power supply voltage Vs, a voltage applied to the first arm  31 , the bus voltage Vdc, and a voltage applied across the ends of the switching element to control the switching element  321  and the switching element  322 . In the case of estimating that a current flows in a body diode of a switching element by using any of the power supply voltage Vs, the voltage applied to the first arm  31 , and the bus voltage Vdc, there are many factors of variation in determination and thus attention should be given to estimation error. In addition, in the case of estimating that a current flows in a body diode of a switching element by using the voltage applied across the switching element, a voltage detecting circuit is required for each of the switching elements for which a current flow is to be estimated. 
     While the example in which the synchronous rectification control is performed by detecting the power supply current Is is described in the first embodiment, the power converting apparatus  100  according to the first embodiment may have a configuration to perform synchronous rectification control by detecting a current flowing through a bus between the bridge circuit  3  and the smoothing capacitor  4  instead of the power supply current Is. In this case, because the current in a short-circuit path cannot be detected, the synchronous rectification control using a current threshold may shorten the period during which the synchronous rectification operation can be performed. Thus, in the case of performing the synchronous rectification control by detecting a bus current, control may be performed such that the switching element  321  or the switching element  322  is turned ON depending on the polarity even when the absolute value of the power supply current Is is smaller than the threshold during the operation with the short-circuit current as described above. In this case, the synchronous rectification operation can be performed for a long period; therefore, the conduction loss of the switching element  321  or the switching element  322  can be reduced. 
     Note that it is desirable that the first arm  31  be configured as a so-called 2-in-1 module in which the switching elements  311  and  312  are provided in one package. Similarly, it is desirable that the second arm  32  be configured as a 2-in-1 module in which the switching elements  321  and  322  are provided in one package. In a 2-in-1 module, two switching elements having the same switching characteristics are often mounted. When each of the first arm  31  and the second arm  32  is configured as a 2-in-1 module, the imbalance in heat generation between the switching element  311  and the switching element  312  is reduced and further, the imbalance in heat generation between the switching element  321  and the switching element  322  is reduced, as compared with the case where the switching elements  311 ,  312 ,  321 , and  322  are each configured as one module. 
     As described above, according to the first embodiment, because an increase in the power supply voltage for the driving circuit  311 DC can be prevented or reduced, a decrease in the withstand voltage of the driving circuit can be prevented or reduced, a decrease in the short circuit withstand of the switching element can be prevented or reduced, and further, the imbalance in heat generation between the switching element  311  and the switching element  312  can be reduced. As a result, the reliability of the power converting apparatus  100  can be improved. In addition, because an increase in the power supply voltage for the driving circuit  311 DC can be prevented or reduced, an isolated power supply for improving a dielectric strength need not be additionally provided between the bootstrap circuit  401  and the driving circuit  311 DC, the structure of the power converting apparatus  100  is simplified, and the manufacturing cost of the power converting apparatus  100  can be reduced. In addition, because the voltage of the direct-current voltage source  300  need not be reduced in order to prevent or reduce an increase in the capacitor voltage, the power supply voltage for the driving circuit  311 DC can be adjusted to a value equal to the power supply voltage for the driving circuit  312 DC while the power supply voltage with which the driving circuit  312 DC can operate is ensured. Consequently, the imbalance in heat generation between the switching element  311  and the switching element  312  can be reduced, and the reliability of the power converting apparatus  100  is improved. In addition, because the power supply voltage for the driving circuit  311 DC can be adjusted to a value equal to the power supply voltage for the driving circuit  312 DC, the loss caused when one of the power supply voltages becomes higher than necessary during switching operations is reduced, the power consumption of the power converting apparatus  100  is reduced, and the efficiency of the power converting apparatus  100  can be improved. In addition, because an increase in the power supply voltage for the driving circuit  311 DC can be prevented or reduced even when switching elements, such as WBG MOSFETs, with inferior forward current-forward voltage characteristics of the body diodes are used, the first embodiment is suitable for the power converting apparatus  100  including WBG MOSFETs, in particular, SiC MOSFETs. In addition, the first embodiment is suitable for the power converting apparatus  100  including switching elements, such as WBG switching elements, having characteristics sensitive to a gate driving voltage. 
     The sensitivity to the gate driving voltage will be explained. A conduction loss and a switching loss are used for performance indices of a SiC MOSFET. The conduction loss is determined by the ON-resistance and the current value of a MOSFET, and the ON-resistance is known to vary significantly depending on the gate driving voltage. Typically, the ON-resistance exhibits a tendency to rapidly increase when the gate driving voltage is low, and converges to a specific value as the gate driving voltage becomes higher. A semiconductor, however, has an element withstand voltage; therefore, the gate driving voltage cannot be increased unlimitedly. In such a case where the ON-resistance converges to a specific value when the gate driving voltage is 16 to 18 V, for example, the ON-resistance becomes twice the specific value when the gate driving voltage is reduced to 10 V. The change in the ON-resistance depending on the value of the gate driving voltage in this manner is referred to as sensitivity to the gate driving voltage in the present embodiment. 
     Second Embodiment 
     While a switching element pair constituted by two switching elements connected in series is provided in the first arm  31  in the first embodiment, a configuration in which n pairs of switching elements are connected in parallel in the first arm  31  and synchronous control is performed thereon will be described in a second embodiment. Here, n is an integer not smaller than 2.  FIG. 30  is a diagram illustrating an example of a configuration of a power converting apparatus according to the second embodiment. In a power converting apparatus  100 A according to the second embodiment, the first arm  31  includes a switching element  313  that is a fifth switching element and a switching element  314  that is a sixth switching element. The switching element  313  and the switching element  314  are connected in series. A switching element pair constituted by the switching element  313  and the switching element  314  is connected in parallel with the switching element pair constituted by the switching element  311  and the switching element  312 . A reactor  2  is connected at a connection point of the switching element  313  and the switching element  314 .  FIG. 30  illustrates an example of a configuration in which synchronous control is performed by using two arms. 
     When driving the first arm  31  in which two pairs of switching elements are connected in parallel, the control unit  10  drives two switching elements  311  and  313  constituting an upper arm simultaneously and drives two switching elements  312  and  314  constituting a lower arm simultaneously, among the two pairs of switching elements. Note that simultaneously driving two switching elements connected in parallel with each other will be referred to as “parallel driving”. 
     The parallel driving of two pairs of switching elements connected in parallel reduces the current flowing in each of the switching elements to half of that in the case of one pair of switching elements. As is clear from the characteristics in  FIG. 19 , as the current is smaller, the loss of the switching element is smaller, and the loss occurring in the first arm  31  is thus reduced. Consequently, the imbalance in heat generation between the first arm  31  and the second arm  32  can further be reduced. 
     While an example of the configuration in which two pairs of switching elements are connected in parallel is illustrated in  FIG. 30 , the number of pairs of switching elements is not limited to two, and may be n. In a case where the first arm  31  is constituted by n pairs of switching elements, the current flowing in one pair of switching elements is reduced to one n-th, and the loss in the first arm  31  can thus be further reduced. Note that the imbalance in loss among n pairs of switching elements connected in parallel need not be completely eliminated, and the number of pairs of switching elements to be connected in parallel may be selected within a range in which the imbalance in loss is permitted. 
     In addition, simultaneous driving of two switching elements connected in parallel in the first arm  31  is explained in the example of  FIG. 30 . Thus, in the second embodiment, a synchronous control method of simultaneously switching the switching elements connected in parallel is employed. The method of controlling the switching elements connected in parallel, however, is not limited thereto, and so-called interleaved control in which the phases of two switching elements connected in parallel are shifted by 180° from each other for control may be used. 
     In the interleaved control, the phases when the switching element  311  and the switching element  313  connected in parallel are turned ON are shifted by 180° from each other for control, and the phases when the switching element  312  and the switching element  314  connected in parallel are turned on are shifted by 180° from each other for control. As a result, the two switching elements connected in parallel are subjected to interleaved driving. 
     Interleaved driving of the first arm  31  facilitates driving at higher frequency, and enables reduction in size of the reactor  2  and reduction in the reactor loss. Note that, in a case of being frequently used in the passive state like the case with conditioners, the reactor  2  need not be reduced in size, and the configurations and operations of the first embodiment are more effective in terms of harmonic wave prevention and the power-supply power factor. 
     While one reactor  2  is provided between the alternating-current power supply  1  and the first arm  31  in the first and second embodiments, the configurations of the first and second embodiments are not limited thereto, and a reactor may also be provided between the alternating-current power supply  1  and the second arm  32 . Use of two reactors in this manner can make the capacity of each reactor smaller, which improves the design freedom of the power converting apparatuses  100  and  100 A as compared with a case where one reactor with a large capacity is used. 
     A hardware configuration of the control unit  10  of the power converting apparatuses  100  and  100 A according to the first and second embodiments will now be described.  FIG. 31  is a diagram illustrating an example of the hardware configuration for implementing the control unit of the first and second embodiments. The control unit  10  described in the first and second embodiments is implemented by a processor  201  and a memory  202 . 
     The processor  201  is a central processing unit (CPU; also referred to as a central processing device, a processing device, a computing device, a microprocessor, a microcomputer, a processor, or a digital signal processor (DSP)), or a system large scale integration (LSI). The memory  202  is a semiconductor memory such as a random access memory (RAM), a read only memory (ROM), a flash memory, an erasable programmable read only memory (EPROM), or an electrically erasable programmable read only memory (EEPROM: registered trademark). The semiconductor memory may be a nonvolatile memory or may be a volatile memory. Alternatively, the memory  202  may be a magnetic disk, a flexible disk, an optical disk, a compact disk, a mini disc, or a digital versatile disc (DVD) instead of a semiconductor memory. 
     The power supply current command value control unit  21 , the ON-duty control unit  22 , the power supply voltage phase calculating unit  23 , the first pulse generating unit  24 , the second pulse generating unit  25 , the current command value calculating unit  26 , and the instantaneous value command value calculating unit  27  illustrated in  FIG. 13  are implemented by the processor  201  and the memory  202  illustrated in  FIG. 31 . Specifically, the respective units are implemented by the processor  201  by storing programs for operating as each of the power supply current command value control unit  21 , the ON-duty control unit  22 , the power supply voltage phase calculating unit  23 , the first pulse generating unit  24 , the second pulse generating unit  25 , the current command value calculating unit  26 , and the instantaneous value command value calculating unit  27  in the memory  202  and reading and executing the programs stored in the memory  202 . 
     Third Embodiment 
       FIG. 32  is a diagram illustrating an example of a configuration of a motor driving apparatus according to a third embodiment. A motor driving apparatus  101  according to a third embodiment drives a motor  42  that is a load. The motor driving apparatus  101  includes the power converting apparatus  100  of the first embodiment, an inverter  41 , a motor current detecting unit  44 , and an inverter controlling unit  43 . The inverter  41  drives the motor  42  by converting a direct-current power supplied from the power converting apparatus  100  into an alternating-current power and outputting the alternating-current power to the motor  42 . 
     Note that the motor driving apparatus  101  may include the power converting apparatus  100 A of the second embodiment instead of the power converting apparatus  100  of the first embodiment. In addition, while the load of the motor driving apparatus  101 , that is, the device connected to the inverter  41  is the motor  42  in the third embodiment, the device connected to the inverter  41  may be any device, other than the motor  42 , to which an alternating-current power is input. 
     The inverter  41  is a circuit including switching elements, including insulated gate bipolar transistors (IGBTs), in a three-phase bridge configuration or a two-phase bridge configuration. The switching elements included in the inverter  41  are not limited to IGBTs, but may be switching elements made of WBG semiconductors, insulated gate controlled thyristors (IGCTs), field effect transistors (FETs) or MOSFETs. 
     The motor current detecting unit  44  detects a current flowing between the inverter  41  and the motor  42 . The inverter controlling unit  43  generates PWM signals for driving the switching elements in the inverter  41  by using a current detected by the motor current detecting unit  44  such that the motor  42  rotates at a rotating speed, and outputs the generated PWM signals to the inverter  41 . The inverter controlling unit  43  is implemented by a processor and a memory in a manner similar to the control unit  10 . Note that the inverter controlling unit  43  of the motor driving apparatus  101  and the control unit  10  of the power converting apparatus  100  may be implemented by one circuit. 
     In a case where the power converting apparatus  100  or  100 A according to the first or second embodiment is used in the motor driving apparatus  101 , the bus voltage Vdc necessary for controlling the bridge circuit  3  illustrated in  FIG. 1  and  FIG. 30  changes depending on the operation state of the motor  42 . Typically, as the rotating speed of the motor  42  is higher, the voltage output from the inverter  41  need to be higher. The upper limit of the voltage output from the inverter  41  is limited by a voltage input to the inverter  41 , that is, the bus voltage Vdc that is output from the power converting apparatus  100  or  100 A. A region in which the voltage output from the inverter  41  exceeds the upper limit limited by the bus voltage Vdc and saturated is called an overmodulation region. 
     In the motor driving apparatus  101  as described above, the bus voltage Vdc need not be increased in a low rotation range of the motor  42 , that is, in a range in which the overmodulation region is not reached. In contrast, when the motor  42  rotates at high speed, the overmodulation region can be shifted toward higher rotation by increasing the bus voltage Vdc. As a result, the operation range of the motor  42  can be expanded toward higher rotation. 
     In addition, when the operation range of the motor  42  need not be expanded, the number of coil turns around a stator of the motor  42  can be increased by a corresponding amount. In the low rotation region, the increase in the number of coil turns makes the motor voltage generated across the coil ends higher and lowers the current flowing in the coil accordingly, which reduces the loss caused by the switching operation of the switching elements in the inverter  41 . For producing both effects of expansion of the operation range of the motor  42  and improvement in the loss in the low rotation region, the number of coil turns of the motor  42  is set to an appropriate value. 
     According to the third embodiment, because the power converting apparatus  100  or  100 A according to the first or second embodiment is used, the effect of improving the reliability of the motor driving apparatus  101  is produced. In addition, because an increase in the temperature of the motor driving apparatus  101  is prevented or reduced as a result of applying switching elements made of WBG semiconductors to the power converting apparatus  100  or  100 A according to the first or second embodiment, the capacity of cooling the components mounted on the motor driving apparatus  101  can be ensured even when the motor driving apparatus  101  is reduced in size. In addition, high-frequency driving of switching elements made of WBG semiconductors enables reduction in size and loss of the reactor  2 . Thus, as a result of applying switching elements made of WBG semiconductors to the power converting apparatus  100  or  100 A according to the first or second embodiment, an increase in weight of the motor driving apparatus  101  can be prevented or reduced. 
     Fourth Embodiment 
       FIG. 33  is a diagram illustrating an example of a configuration of an air conditioner according to a fourth embodiment. An air conditioner  700  according to the fourth embodiment is an example of a refrigeration cycle system, and includes the motor driving apparatus  101  according to the third embodiment, and a motor  42 . The air conditioner  700  also includes a compressor  81 , a four-way valve  82 , an outdoor heat exchanger  83 , an expansion valve  84 , an indoor heat exchanger  85 , and refrigerant piping  86 . 
     The air conditioner  700  may be a split air conditioner in which an outdoor unit is separated from an indoor unit, or may be an integrated air conditioner in which the compressor  81 , the indoor heat exchanger  85 , and the outdoor heat exchanger  83  are arranged in one housing. 
     The compressor  81  includes therein a compression mechanism  87  for compressing the refrigerant, and a motor  42  for causing the compression mechanism  87  to operate. The motor  42  is driven by the motor driving apparatus  101 . In the air conditioner  700 , a refrigeration cycle is constituted by circulation of refrigerant through the compressor  81 , the four-way valve  82 , the outdoor heat exchanger  83 , the expansion valve  84 , the indoor heat exchanger  85 , and the refrigerant piping  86 . 
     Note that the components of the air conditioner  700  can also be applied to such equipment as a refrigerator or a freezer including a refrigeration cycle. In addition, while the motor  42  is used for a driving source of the compressor  81  in the fourth embodiment, the motor  42  may be used as a driving source for driving each of an indoor unit fan and an outdoor unit fan, which are not illustrated, instead of the compressor  81 . Alternatively, the motor  42  may be applied to a driving source for each of the indoor unit fan, the outdoor unit fan, and the compressor  81 , and the three motors  42  may be driven by the motor driving apparatus  101 . 
     In addition, because the operation of the air conditioner  700  under an intermediate condition in which the power output is equal to or lower than half of a rated power output, that is the operation of the air conditioner  700  in a low power output range is dominant throughout the year, the contribution to the annual power consumption under the intermediate condition is high. In addition, in the air conditioner  700 , the rotating speed of the motor  42  tends to be low, and the bus voltage required for driving the motor  42  tends to be low. Thus, operation of the switching elements used in the air conditioner  700  in a passive state is effective in terms of system efficiency. The power converting apparatus  100  capable of reducing the loss in a wide range of operation modes from the passive state to the high-frequency switching state is therefore useful for the air conditioner  700 . Although the reactor  2  can be reduced in size with the interleaved control as described above, the frequency of operation of the air conditioner  700  under the intermediate condition is high and thus the reactor  2  need not be reduced in size; therefore, the configurations and operations of the power converting apparatus  100  or  100 A according to the first or second embodiment are more effective in terms of harmonic wave prevention and the power-supply power factor. 
     In addition, as described above, because the switching loss in the case where switching elements made of WBG semiconductors are driven at a high switching frequency equal to or higher than 10 kHz is smaller than that of switching elements made of Si semiconductors, application of switching elements made of WBG semiconductors to the power converting apparatus  100  or  100 A according to the first or second embodiment prevents or reduces an increase in temperature of the motor driving apparatus  101 . As a result, the capacity of cooling the components mounted on the motor driving apparatus  101  can be ensured even when the outdoor unit fan is reduced in size. The power converting apparatus  100  or  100 A according to the first or second embodiment is therefore suitable for use in the air conditioner  700  that is highly efficient and has a high power equal to or higher than 4.0 kW. 
     In addition, switching elements made of WBG semiconductors can be driven at higher frequency than switching elements made of Si semiconductors. Thus, high-frequency driving enables reduction in size and in loss of the reactor  2 . Thus, as a result of applying switching elements made of WBG semiconductors to the power converting apparatus  100  or  100 A according to the first or second embodiment, an increase in weight of the air conditioner  700  can be prevented or reduced. 
     In addition, according to the fourth embodiment, high-frequency driving of the switching elements reduces the switching loss, and the air conditioner  700  with a low energy consumption rate and high efficiency can thus be achieved. 
     The configurations presented in the embodiments above are examples of the present invention, and can be combined with other known technologies or can be partly omitted or modified without departing from the scope of the present invention. 
     REFERENCE SIGNS LIST 
       1  single-phase alternating-current power supply;  2  reactor;  3  bridge circuit;  4  smoothing capacitor;  5  power supply voltage detecting unit;  6  power supply current detecting unit;  7  bus voltage detecting unit;  10  control unit;  11 ,  311 ,  312 ,  313 ,  314 ,  321 ,  322  switching element;  21  power supply current command value control unit;  22  on-duty control unit;  23  power supply voltage phase calculating unit;  24  first pulse generating unit;  25  second pulse generating unit;  26  current command value calculating unit;  27  instantaneous value command value calculating unit;  31  first arm;  32  second arm;  41  inverter;  42  motor;  43  inverter controlling unit;  44  motor current detecting unit;  50  load;  81  compressor;  82  four-way valve;  83  outdoor heat exchanger;  84  expansion valve;  85  indoor heat exchanger;  86  refrigerant piping;  87  compression mechanism;  100 ,  100 - 1 ,  100 - 2 ,  100 - 3 ,  100 A power converting apparatus;  101  motor driving apparatus;  201  processor;  202  memory;  241  carrier generating unit;  242  reference PWM generating unit;  243  dead time generating unit;  244  pulse selector;  300  direct-current voltage source;  311 C,  321 C boot capacitor;  311 D,  321 D boot diode;  311 D′,  312 RD,  321 D′,  322 RD gate voltage suppression diode;  311 DC,  312 DC,  321 DC,  322 DC driving circuit;  311 R,  321 R boot resistor;  311   a ,  312   a ,  321   a ,  322   a  body diode;  312 BD body diode voltage;  401 ,  401 A,  402 ,  402 A bootstrap circuit;  501  first line;  502  second line;  503  third line;  504  fourth line;  506  first connection point;  508  second connection point;  600  semiconductor substrate;  601 ,  603  region;  602  insulating oxide layer;  604  channel;  700  air conditioner.