Patent Publication Number: US-8121323-B2

Title: Inter-channel communication in a multi-channel digital hearing instrument

Description:
This is a continuation of U.S. patent application Ser. No. 10/125,184, file Apr. 18, 2002, now U.S. Pat. No. 7,181,034, which claims priority from and is related to the following prior application: Inter-Channel Communication In a Multi-Channel Digital Hearing Instrument, U.S. Provisional Application No. 60/284,459, filed Apr. 18, 2001. 
    
    
     BACKGROUND 
     1. Field of the Invention 
     This invention generally relates to digital hearing aid instruments. More specifically, the invention provides an advanced inter-channel communication system and method for multi-channel digital hearing aid instruments. 
     2. Description of the Related Art 
     Digital hearing aid instruments are known in this field. Multi-channel digital hearing aid instruments split the wide-bandwidth audio input signal into a plurality of narrow-bandwidth sub-bands, which are then digitally processed by an on-board digital processor in the instrument. In first generation multi-channel digital hearing aid instruments, each sub-band channel was processed independently from the other channels. Subsequently, some multi-channel instruments provided for coupling between the sub-band processors in order to refine the multi-channel processing to account for masking from the high-frequency channels down towards the lower-frequency channels. 
     A low frequency tone can sometimes mask the user&#39;s ability to hear a higher frequency tone, particularly in persons with hearing impairments. By coupling information from the high-frequency channels down towards the lower frequency channels, the lower frequency channels can be effectively turned down in the presence of a high frequency component in the signal, thus unmasking the high frequency tone. The coupling between the sub-bands in these instruments, however, was uniform from sub-band to sub-band, and did not provide for customized coupling between any two of the plurality of sub-bands. In addition, the coupling in these multi-channel instruments did not take into account the overall content of the input signal. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of an exemplary digital hearing aid system according to the present invention. 
         FIG. 2  is an expanded block diagram of the channel processing/twin detector circuitry shown in  FIG. 1 . 
         FIG. 3  is an expanded block diagram of one of the mixers shown in  FIG. 2 . 
     
    
    
     SUMMARY 
     A multi-channel digital hearing instrument is provided that includes a microphone, an analog-to-digital (A/D) converter, a sound processor, a digital-to-analog (D/A) converter and a speaker. The microphone receives an acoustical signal and generates an analog audio signal. The A/D converter converts the analog audio signal into a digital audio signal. The sound processor includes channel processing circuitry that filters the digital audio signal into a plurality of frequency band-limited audio signals and that provides an automatic gain control function that permits quieter sounds to be amplified at a higher gain than louder sounds and may be configured to the dynamic hearing range of a particular hearing instrument user. The D/A converter converts the output from the sound processor into an analog audio output signal. The speaker converts the analog audio output signal into an acoustical output signal that is directed into the ear canal of the hearing instrument user. 
     DETAILED DESCRIPTION 
     Turning now to the drawing figures,  FIG. 1  is a block diagram of an exemplary digital hearing aid system  12 . The digital hearing aid system  12  includes several external components  14 ,  16 ,  18 ,  20 ,  22 ,  24 ,  26 ,  28 , and, preferably, a single integrated circuit (IC)  12 A. The external components include a pair of microphones  24 ,  26 , a tele-coil  28 , a volume control potentiometer  24 , a memory-select toggle switch  16 , battery terminals  18 ,  22 , and a speaker  20 . 
     Sound is received by the pair of microphones  24 ,  26 , and converted into electrical signals that are coupled to the FMIC  12 C and RMIC  12 D inputs to the IC  12 A. FMIC refers to “front microphone,” and RMIC refers to “rear microphone.” The microphones  24 ,  26  are biased between a regulated voltage output from the RREG and FREG pins  12 B, and the ground nodes FGND  12 F and RGND  12 G. The regulated voltage output on FREG and RREG is generated internally to the IC  12 A by regulator  30 . 
     The tele-coil  28  is a device used in a hearing aid that magnetically couples to a telephone handset and produces an input current that is proportional to the telephone signal. This input current from the tele-coil  28  is coupled into the rear microphone A/D converter  32 B on the IC  12 A when the switch  76  is connected to the “T” input pin  12 E, indicating that the user of the hearing aid is talking on a telephone. The tele-coil  28  is used to prevent acoustic feedback into the system when talking on the telephone. 
     The volume control potentiometer  14  is coupled to the volume control input  12 N of the IC. This variable resistor is used to set the volume sensitivity of the digital hearing aid. 
     The memory-select toggle switch  16  is coupled between the positive voltage supply VB  18  and the memory-select input pin  12 L. This switch  16  is used to toggle the digital hearing aid system  12  between a series of setup configurations. For example, the device may have been previously programmed for a variety of environmental settings, such as quiet listening, listening to music, a noisy setting, etc. For each of these settings, the system parameters of the IC  12 A may have been optimally configured for the particular user. By repeatedly pressing the toggle switch  16 , the user may then toggle through the various configurations stored in the read-only memory  44  of the IC  12 A. 
     The battery terminals  12 K,  12 H of the IC  12 A are preferably coupled to a single 1.3 volt zinc-air battery. This battery provides the primary power source for the digital hearing aid system. 
     The last external component is the speaker  20 . This element is coupled to the differential outputs at pins  12 J,  12 I of the IC  12 A, and converts the processed digital input signals from the two microphones  24 ,  26  into an audible signal for the user of the digital hearing aid system  12 . 
     There are many circuit blocks within the IC  12 A. Primary sound processing within the system is carried out by a sound processor  38  and a directional processor and headroom expander  50 . A pair of A/D converters  32 A,  32 B are coupled between the front and rear microphones  24 ,  26 , and the directional processor and headroom expander  50 , and convert the analog input signals into the digital domain for digital processing. A single D/A converter  48  converts the processed digital signals back into the analog domain for output by the speaker  20 . Other system elements include a regulator  30 , a volume control A/D  40 , an interface/system controller  42 , an EEPROM memory  44 , a power-on reset circuit  46 , a oscillator/system clock  36 , a summer  71 , and an interpolator and peak clipping circuit  70 . 
     The sound processor  38  preferably includes a pre-filter  52 , a wide-band twin detector  54 , a band-split filter  56 , a plurality of narrow-band channel processing and twin detectors  58 A- 58 D, a summation block  60 , a post filter  62 , a notch filter  64 , a volume control circuit  66 , an automatic gain control output circuit  68 , an interpolator and peak clipping circuit  70 , a squelch circuit  72 , a summation block  71 , and a tone generator  74 . 
     Operationally, the digital hearing aid system  12  processes digital sound as follows. Analog audio signals picked up by the front and rear microphones  24 ,  26  are coupled to the front and rear A/D converters  32 A,  32 B, which are preferably Sigma-Delta modulators followed by decimation filters that convert the analog audio inputs from the two microphones into equivalent digital audio signals. Note that when a user of the digital hearing aid system is talking on the telephone, the rear A/D converter  32 B is coupled to the tele-coil input “T”  12 E via switch  76 . Both the front and rear A/D converters  32 A,  32 B are clocked with the output clock signal from the oscillator/system clock  36  (discussed in more detail below). This same output clock signal is also coupled to the sound processor  38  and the D/A converter  48 . 
     The front and rear digital sound signals from the two A/D converters  32 A,  32 B are coupled to the directional processor and headroom expander  50  of the sound processor  38 . The rear A/D converter  32 B is coupled to the processor  50  through switch  75 . In a first position, the switch  75  couples the digital output of the rear A/D converter  32  B to the processor  50 , and in a second position, the switch  75  couples the digital output of the rear A/D converter  32 B to summation block  71  for the purpose of compensating for occlusion. 
     Occlusion is the amplification of the users own voice within the ear canal. The rear microphone can be moved inside the ear canal to receive this unwanted signal created by the occlusion effect. The occlusion effect is usually reduced by putting a mechanical vent in the hearing aid. This vent, however, can cause an oscillation problem as the speaker signal feeds back to the microphone(s) through the vent aperture. Another problem associated with traditional venting is a reduced low frequency response (leading to reduced sound quality). Yet another limitation occurs when the direct coupling of ambient sounds results in poor directional performance, particularly in the low frequencies. The system shown in  FIG. 1  solves these problems by canceling the unwanted signal received by the rear microphone  26  by feeding back the rear signal from the A/D converter  32 B to summation circuit  71 . The summation circuit  71  then subtracts the unwanted signal from the processed composite signal to thereby compensate for the occlusion effect. 
     The directional processor and headroom expander  50  includes a combination of filtering and delay elements that, when applied to the two digital input signals, form a single, directionally-sensitive response. This directionally-sensitive response is generated such that the gain of the directional processor  50  will be a maximum value for sounds coming from the front microphone  24  and will be a minimum value for sounds coming from the rear microphone  26 . 
     The headroom expander portion of the processor  50  significantly extends the dynamic range of the A/D conversion, which is very important for high fidelity audio signal processing. It does this by dynamically adjusting the operating points of the A/D converters  32 A/ 32 B. The headroom expander  50  adjusts the gain before and after the A/D conversion so that the total gain remains unchanged, but the intrinsic dynamic range of the A/D converter block  32 A/ 32 B is optimized to the level of the signal being processed. 
     The output from the directional processor and headroom expander  50  is coupled to the pre-filter  52  in the sound processor, which is a general-purpose filter for pre-conditioning the sound signal prior to any further signal processing steps. This “pre-conditioning” can take many forms, and, in combination with corresponding “post-conditioning” in the post filter  62 , can be used to generate special effects that may be suited to only a particular class of users. For example, the pre-filter  52  could be configured to mimic the transfer function of the user&#39;s middle ear, effectively putting the sound signal into the “cochlear domain.” Signal processing algorithms to correct a hearing impairment based on, for example, inner hair cell loss and outer hair cell loss, could be applied by the sound processor  38 . Subsequently, the post-filter  62  could be configured with the inverse response of the pre-filter  52  in order to convert the sound signal back into the “acoustic domain” from the “cochlear domain.” Of course, other pre-conditioning/post-conditioning configurations and corresponding signal processing algorithms could be utilized. 
     The pre-conditioned digital sound signal is then coupled to the band-split filter  56 , which preferably includes a bank of filters with variable corner frequencies and pass-band gains. These filters are used to split the single input signal into four distinct frequency bands. The four output signals from the band-split filter  56  are preferably in-phase so that when they are summed together in summation block  60 , after channel processing, nulls or peaks in the composite signal (from the summation block) are minimized. 
     Channel processing of the four distinct frequency bands from the band-split filter  56  is accomplished by a plurality of channel processing/twin detector blocks  58 A- 58 D. Although four blocks are shown in  FIG. 1 , it should be clear that more than four (or less than four) frequency bands could be generated in the band-split filter  56 , and thus more or less than four channel processing/twin detector blocks  58  may be utilized with the system. 
     Each of the channel processing/twin detectors  58 A- 58 D provide an automatic gain control (“AGC”) function that provides compression and gain on the particular frequency band (channel) being processed. Compression of the channel signals permits quieter sounds to be amplified at a higher gain than louder sounds, for which the gain is compressed. In this manner, the user of the system can hear the full range of sounds since the circuits  58 A- 58 D compress the full range of normal hearing into the reduced dynamic range of the individual user as a function of the individual user&#39;s hearing loss within the particular frequency band of the channel. 
     The channel processing blocks  58 A- 58 D can be configured to employ a twin detector average detection scheme while compressing the input signals. This twin detection scheme includes both slow and fast attack/release tracking modules that allow for fast response to transients (in the fast tracking module), while preventing annoying pumping of the input signal (in the slow tracking module) that only a fast time constant would produce. The outputs of the fast and slow tracking modules are compared, and the compression parameters are then adjusted accordingly. For example, if the output level of the fast tracking module exceeds the output level of the slow tracking module by some pre-selected level, such as 6 dB, then the output of the fast tracking module may be temporarily coupled as the input to a gain calculation block (see  FIG. 3 ). The compression ratio, channel gain, lower and upper thresholds (return to linear point), and the fast and slow time constants (of the fast and slow tracking modules) can be independently programmed and saved in memory  44  for each of the plurality of channel processing blocks  58 A- 58 D. 
       FIG. 1  also shows a communication bus  59 , which may include one or more connections for coupling the plurality of channel processing blocks  58 A- 58 D. This inter-channel communication bus  59  can be used to communicate information between the plurality of channel processing blocks  58 A- 58 D such that each channel (frequency band) can take into account the “energy” level (or some other measure) from the other channel processing blocks. Preferably, each channel processing block  58 A- 58 D would take into account the “energy” level from the higher frequency channels. In addition, the “energy” level from the wide-band detector  54  may be used by each of the relatively narrow-band channel processing blocks  58 A- 58 D when processing their individual input signals. 
     After channel processing is complete, the four channel signals are summed by summation bock  60  to form a composite signal. This composite signal is then coupled to the post-filter  62 , which may apply a post-processing filter function as discussed above. Following post-processing, the composite signal is then applied to a notch-filter  64 , that attenuates a narrow band of frequencies that is adjustable in the frequency range where hearing aids tend to oscillate. This notch filter  64  is used to reduce feedback and prevent unwanted “whistling” of the device. Preferably, the notch filter  64  may include a dynamic transfer function that changes the depth of the notch based upon the magnitude of the input signal. 
     Following the notch filter  64 , the composite signal is coupled to a volume control circuit  66 . The volume control circuit  66  receives a digital value from the volume control A/D  40 , which indicates the desired volume level set by the user via potentiometer  14 , and uses this stored digital value to set the gain of an included amplifier circuit. 
     From the volume control circuit, the composite signal is coupled to the AGC-output block  68 . The AGC-output circuit  68  is a high compression ratio, low distortion limiter that is used to prevent pathological signals from causing large scale distorted output signals from the speaker  20  that could be painful and annoying to the user of the device. The composite signal is coupled from the AGC-output circuit  68  to a squelch circuit  72 , that performs an expansion on low-level signals below an adjustable threshold. The squelch circuit  72  uses an output signal from the wide-band detector  54  for this purpose. The expansion of the low-level signals attenuates noise from the microphones and other circuits when the input S/N ratio is small, thus producing a lower noise signal during quiet situations. Also shown coupled to the squelch circuit  72  is a tone generator block  74 , which is included for calibration and testing of the system. 
     The output of the squelch circuit  72  is coupled to one input of summation block  71 . The other input to the summation bock  71  is from the output of the rear A/D converter  32 B, when the switch  75  is in the second position. These two signals are summed in summation block  71 , and passed along to the interpolator and peak clipping circuit  70 . This circuit  70  also operates on pathological signals, but it operates almost instantaneously to large peak signals and is high distortion limiting. The interpolator shifts the signal up in frequency as part of the D/A process and then the signal is clipped so that the distortion products do not alias back into the baseband frequency range. 
     The output of the interpolator and peak clipping circuit  70  is coupled from the sound processor  38  to the D/A H-Bridge  48 . This circuit  48  converts the digital representation of the input sound signals to a pulse density modulated representation with complimentary outputs. These outputs are coupled off-chip through outputs  12 J,  12 I to the speaker  20 , which low-pass filters the outputs and produces an acoustic analog of the output signals. The D/A H-Bridge  48  includes an interpolator, a digital Delta-Sigma modulator, and an H-Bridge output stage. The D/A H-Bridge  48  is also coupled to and receives the clock signal from the oscillator/system clock  36  (described below). 
     The interface/system controller  42  is coupled between a serial data interface pin  12 M on the IC  12 , and the sound processor  38 . This interface is used to communicate with an external controller for the purpose of setting the parameters of the system. These parameters can be stored on-chip in the EEPROM  44 . If a “black-out” or “brown-out” condition occurs, then the power-on reset circuit  46  can be used to signal the interface/system controller  42  to configure the system into a known state. Such a condition can occur, for example, if the battery fails. 
       FIG. 2  is an expanded block diagram showing the channel processing/twin detector circuitry  58 A- 58 D shown in  FIG. 1 . This figure also shows the wideband twin detector  54 , the band split filter  56 , which is configured in this embodiment to provide four narrow-bandwidth channels (Ch.  1  through Ch.  4 ), and the summation block  60 . In this figure, it is assumed that Ch.  1  is the lowest frequency channel and Ch.  4  is the highest frequency channel. In this circuit, as described in more detail below, level information from the higher frequency channels are provided down to the lower frequency channels in order to compensate for the masking effect. 
     Each of the channel processing/twin detector blocks  58 A- 58 D include a channel level detector  100 , which is preferably a twin detector as described previously, a mixer circuit  102 , described in more detail below with reference to  FIG. 3 , a gain calculation block  104 , and a multiplier  106 . 
     Each channel (Ch.  1 -Ch.  4 ) is processed by a channel processor/twin detector ( 58 A- 58 D), although information from the wideband detector  54  and, depending on the channel, from a higher frequency channel, is used to determine the correct gain setting for each channel. The highest frequency channel (Ch.  4 ) is preferably processed without information from another narrow-band channel, although in some implementations it could be. 
     Consider, for example, the lowest frequency channel—Ch.  1 . The Ch.  1  output signal from the filter bank  56  is coupled to the channel level detector  100 , and is also coupled to the multiplier  106 . The channel level detector  100  outputs a positive value representative of the RMS energy level of the audio signal on the channel. This RMS energy level is coupled to one input of the mixer  102 . The mixer  102  also receives RMS energy level inputs from a higher frequency channel, in this case from Ch.  2 , and from the wideband detector  54 . The wideband detector  54  provides an RMS energy level for the entire audio signal, as opposed to the level for Ch.  2 , which represents the RMS energy level for the sub-bandwidth associated with this channel. 
     As described in more detail below with reference to  FIG. 3 , the mixer  102  multiplies each of these three RMS energy level inputs by a programmable constant and then combines these multiplied values into a composite level signal that includes information from: (1) the channel being processed; (2) a higher frequency channel; and (3) the wideband level detector. Although  FIG. 2  shows each mixer being coupled to one higher frequency channel, it is possible that the mixer could be coupled to a plurality of higher frequency or lower frequency channels in order to provide a more sophisticated anti-masking scheme. 
     The composite level signal from the mixer is provided to the gain calculation block  104 . The purpose of the gain calculation block  104  is to compute a gain (or volume) level for the channel being processed. This gain level is coupled to the multiplier  106 , which operates like a volume control knob on a stereo to either turn up or down the amplitude of the channel signal output from the filter bank  56 . The outputs from the four channel multipliers  106  are then added by the summation block  60  to form a composite audio output signal. 
     Preferably, the gain calculation block  104  applies an algorithm to the output of the mixer  102  that compresses the mixer output signal above a particular threshold level. In the gain calculation block  104 , the threshold level is subtracted from the mixer output signal to form a remainder. The remainder is then compressed using a log/anti-log operation and a compression multiplier. This compressed remainder is then added back to the threshold level to form the output of the gain processing block  104 . 
       FIG. 3  is an expanded block diagram of one of the mixers  102  shown in  FIG. 2 . The mixer  102  includes three multipliers  110 ,  112 ,  114  and a summation block  116 . The mixer  102  receives three input levels from the wideband detector  54 , the upper channel level, and the channel being processed by the particular mixer  102 . Three, independently-programmable, coefficients C 1 , C 2 , and C 3  are applied to the three input levels by the three multipliers  110 ,  112 , and  114 . The outputs of these multipliers are then added by the summation block  116  to form a composite output level signal. This composite output level signal includes information from the channel being processed, the upper level channel, and from the wideband detector  54 . Thus, the composite output signal is given by the following equation: Composite Level=(Wideband Level*C 3 +Upper Level*C 2 +Channel Level*C 1 ). 
     The technology described herein may provide several advantages over known multi-channel digital hearing instruments. First, the inter-channel processing takes into account information from a wideband detector. This overall loudness information can be used to better compensate for the masking effect. Second, each of the channel mixers includes independently programmable coefficients to apply to the channel levels. This provides for much greater flexibility in customizing the digital hearing instrument to the particular user, and in developing a customized channel coupling strategy. For example, with a four-channel device such as shown in  FIG. 1 , the invention provides for 4,194,304 different settings using the three programmable coefficients on each of the four channels. 
     This written description uses examples to disclose the invention, including the best mode, and also to enable any person skilled in the art to make and use the invention. The patentable scope of the invention is defined by the claims, and may include other examples that occur to those skilled in the art.