Patent Publication Number: US-6670838-B1

Title: Digital clock adaptive duty cycle circuit

Description:
FIELD OF THE INVENTION 
     The invention relates generally to digital circuits that clock transmit data, and more specifically to providing a circuit that dynamically adjusts the waveform of an input digital clock whose nominal duty cycle may not be 50% and to output a digital close having a substantially 50% duty cycle such that transmission data errors are reduced. 
     BACKGROUND OF THE INVENTION 
     In the field of digital communication, it is common to provide a printed circuit board (PCB) whereon several integrated circuits (ICs) are mounted. FIG. 1 shows a generic PCB containing several ICs, IC- 1 , IC- 2 , IC- 3 , and a clock circuit (CLOCK), which itself may be an IC. FIG. 1 depicts the various ICs coupled to receive an operating potential VDD. In many applications, it is desired that the clock frequency be as fast as possible such that DATA can be output quickly. As such, there is a need for ever faster clock frequencies and data rates. 
     FIG. 2A depicts an exemplary waveform for the clock signal (CLOCK) as a function of time. In this example, the logical low voltage magnitude of the clock signal is 0 VDC and the logical high voltage magnitude of the clock is VDD. In FIG. 2A, time T HIGH  denotes the useful high state portion of the clock signal, and T LOW  denotes the useful low state portion of the clock signal. The period of the clock signal is defined as T=T HIGH +T LOW =1/f, where f is the frequency in Hz of the clock signal. The duty cycle (D) of the clock waveform can be defined as D=T HIGH /T or alternatively as D=D LOW /T. The 0-to-1 state portion of the clock waveform defines the rising edge of the clock, and the 1 to-0 state portion of the clock waveform defines the falling edge. In most digital circuits, ICs change state and data is transmitted at the VDD/2 threshold of the rising or falling edge of the clock waveform transitions. As such, increased clock frequency (f) and data transmission between IC&#39;s dictates that the frequency and duty cycle characteristics of the digital signals be more precisely controlled. 
     FIG. 2B depicts an exemplary data signal (DATA), for example an output signal from IC 2  in FIG.  1 . IC 1  may be considered a transmitter IC that provides a CLOCK signal and a DATA signal to IC 2 , which may be considered as a receiver IC. Within IC 2 , the rising and falling edges of the CLOCK signal from IC 1  may be used to “latch” DATA from IC 1  into IC 2 . The time required for the DATA to arrive at IC 2  before the CLOCK signal is present is commonly referred to as the setup time (T SU ). Importantly, if DATA should arrive at IC 2  too early, at a time less than a certain minimum time T SUM , the DATA can be lost, or latched in incorrectly. The amount of time for the DATA to be held after the CLOCK rising edge is commonly referred to as the hold time (T h ). If DATA is removed too soon, e.g., less than a certain time T HM , then again DATA can be lost or latched into IC 2  in error. Thus, T SUM  and T HM  represent the minimum setup time and hold time for error free data transmission. 
     Thus there is a need for a mechanism and method to achieve substantially error free data transmission in a digital circuit that clocks transmit data. Preferably such mechanism and method should ensure that the minimum setup and hold timing requirements are always met, even when the input clock duty cycle is not precisely 50%. 
     The present invention provides such a mechanism and method. 
     SUMMARY OF THE PRESENT INVENTION 
     A digital clock adaptive duty cycle circuit receives an input clock CLKIN having duty cycle of close to 50%, and outputs a CLK signal (and its complement CLKB) whose duty cycle may be continuously and automatically varied to ensure that output duty cycle is 50%, precise to within about ±0.1%. The overall circuit includes a duty cycle adjustor (DCA) unit that includes preferably an odd number of inverter stages. Preferably at least two of the inverter stages include devices, e.g., MOS transistors, that have a parameter (e.g., threshold voltage V TH ) that can be varied as a function of a control voltage V C  to affect the duty cycle of the inverter signal output by the inverter. The effect of V TH  variation within each inverter stage is to vary the duty cycle of the clock signal output from the inverter stage, and thus from the DCA unit itself. The DCA output signal preferably is converted from a single-ended to a differential signal pair, CLK and its complement CLKB. The differential signal pair is low pass filtered and input differentially to an operational amplifier. The output of the operation amplifier is fedback to the DCA unit as control voltage V C . 
     The ability to receive an input clock whose duty cycle may not be exactly 50%, and to dynamically ensure an output CLK signal with a precise 50% duty cycle enables data to be clocked or latch-transferred from IC stage to IC stage substantially error free, even if IC stage setup time varies. The ability to ensure substantially error free data transfer is maintained, even as clock frequency is increased. 
    
    
     Other features and advantages of the invention will appear from the following description in which the preferred embodiments have been set forth in detail, in conjunction with their accompanying drawings. 
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 depicts a generic printed circuit board and clocked integrated circuits, according to the prior art; 
     FIG. 2A depicts and defines regions of a clock signal, according to the prior art; 
     FIG. 2B depicts and defines regions of a data signal, according to the prior art; 
     FIG. 3 depicts a clock circuit with a dynamically adjustable 50% duty cycle, according to the present invention; 
     FIG. 4A depicts a generic CMOS inverter, according to the prior art; 
     FIG. 4B depicts the V IN  VS. V OUT  transfer function for the CMOS inverter of FIG. 4A, according to the prior art; 
     FIG. 5A depicts the CLKIN input clock having 50% duty cycle for an inverter having two inverter threshold levels V TH1  and V TH2 , according to the present invention; 
     FIG. 5B depicts output signal CLK 1  for an inverter having threshold V TH1 &lt;VDD/2 and a resultant less than 50% duty cycle for CLK 1 , according to the present invention; 
     FIG. 5C depicts output signal CLK 2  for an inverter having a threshold V TH2 &gt;VDD/2, and depicts the greater than 50% duty cycle for CLK 2 , according to the present invention; 
     FIGS. 6A and 6B depict two embodiments of a CMOS inverter whose threshold voltage level V TH  can be readily adjusted to alter duty cycle of an inverter output clock signal, according to the present invention; 
     FIG. 6C depicts the transfer function for an inverter such as shown in FIGS. 6A or  6 B, depicting effect of V C  upon V TH  and thus upon inverter output clock signal duty cycle, according to the present invention; 
     FIGS. 7A and 7B depict complementary clock waveforms CLK, CLKB as output by the single-end to differential transformer unit shown in FIG. 3, according to the present invention; and 
     FIG. 8 depicts an exemplary duty cycle adjustor unit comprising one generic and two threshold variable logic inverters, according to the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     As noted with respect to FIGS. 1,  2 A, and  2 B, to achieve substantially error free data transmission requires that at least the minimum setup and hold timing requirements are met. The present invention recognizes that to a first order, T SUM  and T HM  are fixed values that are substantially independent of the clock frequency. Stated differently, regardless of whether the clock frequency is low or high, T SUM  and T HM  must be ensured to attain error free data transmission. 
     A theoretical minimum clock period (T) would allow only for the required time duration T SUM +T HM . Reference is again made to the timing relationship between the clock and data signals shown in FIGS. 2A and 2B, in which T HIGH  and T LOW  for the clock signal, and durations T SU1 , T H1 , T SU2 , and T H2  are shown for two bits of data, denoted BIT  1  and BIT  2 . Minimizing errors during data transmission requires that magnitude of (T SU1 +T H1 ) be substantially equal to magnitude of (T SU2 +T H2 ). Stated differently, minimizing data transmission error requires that the duty cycle of the output clock signal be as close to 50% as possible. As described herein, the present invention receives an input clock signal whose duty cycle nominally is about 50% but in reality may lie in a range of about 33% to about 67%. The present invention then outputs a clock signal whose duty cycle is substantially 50%, e.g., a duty cycle of 50%±1% and more preferably, a duty cycle of 50%±0.1%. 
     When the frequency of the clock increases, the time budgeted to allow for the required T SUM +T HM , decreases. Thus, an imbalance in the clock duty cycle will necessarily impinge on the allotted time for at least one of T SU  and T H . as clock frequency increases, the requirement of a precise 50% clock signal duty cycle becomes more important in terms of reducing data transmission errors. Poor clock duty cycle can result from device mismatches within the clock generator circuitry, which typically is a phase-lock loop (PLL) configuration. Thus in practice, the CLKIN input signal for the present invention may be the output signal from a PLL circuit. (The present invention would not normally be included within the feedback path of a PLL circuit as so doing would add to phase noise associated with the PLL-generated CLKIN clock signal.) 
     The present invention reduces data transmission error by dynamically, e.g., automatically and continuously, adjusting the input clock signal waveform to establish and maintain a substantially 50% duty cycle. FIG. 3 depicts a preferred embodiment of a clock circuit  10  to dynamically attain the desired 50% duty cycle. In FIG. 3, clock circuit  10  receives an input clock signal CLKIN, whose duty cycle is nominally 50% but in practice may range from about 33% to about 67%, and outputs complementary clock signals CLK, CLKB, whose duty cycle is dynamically, i.e., continuously and automatically, adjustable to substantially 50%, according to a first embodiment of the present invention. As such, circuit  10  could be used in the prior art configuration of FIG. 1 in lieu of conventional clock circuit IC- 5 . In practice, the frequency range of the CLK signal output by circuit  10  could be as low as about 1 MHz to as high as perhaps 1,500 MHz. In practice, typical rise and fall times would be on the order of about 0.15 ns. 
     Clock circuit  10  preferably includes a duty cycle adjustor (DCA) unit  20  that receives the CLKIN signal and a feedback signal, a single-ended to differential transformer unit  30 , a low pass filter unit  40 , an output operational amplifier  50 , and a feedback loop  60  that couples the output signal (V C ) from amplifier  50  to DCA unit  20 , which also receives the CLKIN signal, typically a single-ended clock signal. As will be described, output signal V C  is used as a control signal to vary threshold voltage V TH  for inverters comprising DCA unit  20 , and thus to vary duty cycle of the CLK (or CLKB) signal output by circuit  10 . 
     In a preferred embodiment, DCA unit  20  includes a number of series-coupled logic inverters denoted  70 ,  80 ,  90  where an odd number of inverters  90  are disposed between inverters  70  and  80 . In the preferred embodiment, the threshold voltage V TH  associated with each logic inverter  70 ,  80  is adjustable. Exemplary such variable threshold type inverters  70  and  80  are depicted in FIGS. 6A and 6B, described later herein. Inverter(s)  90  may be generic inverter(s) without explicit threshold adjustment capability. 
     It may be useful to first review some basic considerations of logic inverter operation, to better understand the concept of threshold variable logic inverters  70 ,  80  such as used in the present invention. FIG. 4A depicts a generic logic inverter  90 , whose threshold voltage is not per se adjustable. Inverter  90 , and indeed threshold adjustable inverters  70  and  80  may be implemented using CMOS devices such as NMOS device M 1  and PMOS device M 2 , coupled between VDD (or other power source) and ground (or other reference). Referring still to FIG. 4A, a logic high input level for V IN , corresponding here to a high voltage magnitude, will be inverted by inverter  90  to a logic low level, corresponding here to a low voltage magnitude V OUT , at the inverter output and vice versa. Thus, ideally when V IN  is high, V OUT  is low, and when V IN  is low, V OUT  is high. 
     FIG. 4B depicts a typical transfer function for generic inverter  90 . The transition point for inverter  90  is defined when V OUT =V IN , which level is commonly referred as to the threshold level (V TH ) for the inverter. As will be described, variation in inverter threshold voltage V TH  is created to intentionally vary duty cycle of the clock signal output by the inverter. 
     The duty cycle of the V OUT  output signal for inverter  90  is primarily dependent upon two parameters: the duty cycle of the V IN  signal to the inverter, and the magnitude of the inverter threshold V TH . FIG. 5A depicts a generic input clock signal CLKIN with a nominal 50% duty cycle, and also shows two threshold voltage levels: V TH1 &gt;VDD/2, and V TH2 &lt;VDD/2. Assume that CLKIN is input to an inverter, perhaps inverter  90  or more preferably (as will be described with respect to FIGS. 6A and 6B, threshold variable inverters  70  or  80 ). 
     FIG. 5B depicts the inverter output signal, denoted CLK 1 , for the case where the inverter threshold voltage is V TH2 , e.g., a voltage threshold less than the magnitude of VDD/2. According to the present invention, the resultant output duty cycle for the CLK 1  waveform is less than 50%, notwithstanding that the CLKIN input waveform had a nominal 50% duty cycle. 
     By contrast, FIG. 5C depicts the inverter output signal, denoted CLK 2 , for the case where the inverter threshold voltage is V TH1 , e.g., a voltage threshold greater than the magnitude of VDD/2. According to the present invention, the output duty cycle for the CLK 1  waveform is greater than 50%, notwithstanding that the CLKIN input waveform had a nominal 50% duty cycle. 
     It will be appreciated from FIGS. 5A-5C that the duty cycle of an output clock signal, e.g., CLK (or its complement CLKB) in FIG. 3 can be dynamically adjusted by varying the threshold level V TH  for at least one inverter stage within the DCA unit  20  in system  10  shown in FIG.  3 . The duty cycle can be increased or decreased relative to 50% over a range that is determined at least in part by the rise and fall transition times of the input clock signal CLKIN. 
     FIGS. 6A and 6B depict two embodiments of a CMOS inverter  70 ,  80 , whose individual threshold voltage level V TH  can be readily adjusted by varying magnitude of a control voltage V C  The desired result is a controllable variation in the duty cycle between the V IN  signal input to the inverter, and the V OUT  inverted signal output by the inverter. 
     FIG. 6C depicts the transfer-function for a threshold variable inverter such as shown in FIG. 6A or FIG.  6 B. FIG. 6C depicts the inter-relationship between control voltage V C  and inverter threshold voltage V TH . 
     More specifically, FIG. 6A depicts an inverter stage  70  comprising a CMOS inverter (devices M 1 , M 2 ) coupled in series with an NMOS control device M 3 , whose input or gate lead is coupled to receive a control voltage V C . As the magnitude of control voltage V C  decreases, the threshold voltage V TH  for inverter stage  70  will increase, until V TH  approaches the magnitude VDD. In the embodiment of FIG. 6B, inverter stage  80  includes a PMOS device M 4 , whose input or gate lead is coupled to receive control voltage V C . In FIG. 6B, as magnitude of V C  increases, the threshold voltage V TH  for inverter  80  decreases, until V TH  approaches 0 VDC. 
     Referring back to FIG.  3  and referring now to FIG. 8, let duty cycle adjustor unit  20  comprise an odd number (e.g., N=odd integer) of inverters  70 ,  80 , and  90  coupled in series such that the output of one inverter is the input to the next inverter in the series. Varying the magnitude of control voltage V C  advantageously results in adjusting the equivalent threshold voltage V TH  for inverters  70  and  80 , and thus of DCA unit  20 , from 0 VDC to VDD, e.g., the maximum voltage range available, assuming system  10  is powered by a voltage source of magnitude VDD. Note that if inverter  70  and inverter  80  were series-coupled directly to each other, the V C  variation upon V TH  for each inverter would tend to cancel. Thus in practice, an odd number of generic inverters  90  is placed in series between the output of inverter  70  and the input of inverter  80  (or vice-versa if the roles of inverters  70  and  80  are exchanged). If desired, the total number of inverters  70 ,  80 ,  90  could be increased, although in practice a single generic inverter  90  disposed between a single inverter  70  and a single inverter  80  should suffice to implement DCA unit  20 . Such a configuration of inverters within DCA unit  20  will ensure that the effects of V C  upon V TH  and thus upon output signal duty cycle will be additive or subtractive, rather than self-cancelling. 
     Thus in system  10  shown in FIG. 3, duty cycle adjustment unit  20  as shown in FIG. 8 will include inverter(s) such as  70 ,  80 , and  90  to adjust output signal duty cycle up and down, e.g., greater than 50% and less than 50%. The thus-adjusted single-ended intermediate clock signal output from DCA unit  20  is preferably transformed from a single-ended signal to a pair of differential (or complementary) signals, CLK and CLKB, through the single-ended to differential transformation (SE-to-DE XFMR) unit  30 . Unit  30  may be implemented using an inverter and transmission gate in combination (among other implementations). 
     The DC components V CLK , V CLKB  present in the differential clock signal CLK, CLKB output by converter unit  30  are next extracted, for example using a low-pass filter unit  40 . While many filter topologies and characteristics could be used, in one embodiment low-pass filter unit had a cut-off frequency of about 10 MHz and was implemented as a single-pole filter. The DC components extracted by low pass filter unit  40  (or other unit) are input differentially to operational amplifier  50 , whose output signal is coupled via feedback path  60  as a control signal (V C ) to DCA unit  20 . Overall, the cutoff frequency of system  10  is determined primarily by amplifier  50  rather than by low pass filter unit  40 . The cutoff frequency of system  10  is on the order of about 100 KHz, with a bandwidth of about 200 Hz. System  10  could, however, be implemented to exhibit different cutoff and bandwidth frequencies. 
     FIGS. 7A and 7B depict the complementary clock signals CLK, CLKB output by the single-ended to differential ended transformer unit  30  shown in FIG.  3 . Note that the time duration of the CLK high portion of the clock cycle T HIGH  is the same as the CLKB low portion of the clock cycle T′ LOW , and that the time duration of the CLK low portion of the clock cycle T LOW  is the same as the CLKB high portion of the time cycle T′ HIGH . It follows from the above that the duty cycle (D CLK ) of the clock waveform CLK can be related to the duty cycle (D CLKB ) of the complementary clock waveform CLKB as follows, where T is the period of the CLK or CLKB waveform: 
       D   CLK   =T   HIGH   /T=T′   LOW    /T =1 −T′   HIGH   /T =1 −D   CLKB   
     As noted in FIG. 3, the differential clock output waveforms CLK, CLKB from unit  30  are lowpass filtered by unit  40 , which extracts DC levels V CLK , V CLKB  from the CLK and CLKB waveforms. The DC level V CLK  corresponding to CLK can be related to the CLK duty cycle D CLK  as the product V CLK =VDD·D CLK , and the DC level V CLKB  corresponding to CLKB can be related to the CLKB duty cycle D CLKB  as the product  VCLKB =VDD·D CLKB . The V CLK , V CLKB  DC levels are input to high gain operational amplifier  50 , whose output signal is coupled via feedback path  60  as the control signal V C  to duty cycle adjustor  20 , as shown in FIG.  3 . As described earlier herein, the effect of the V C  control signal is to adjust the V TH  thresholds of the inverter stages, e.g.,  80 ,  90 - comprising DCA unit  20 , to adjust the duty cycle of the CLK, and CLKB waveforms. The frequency response of system  10  may be controlled with a capacitor (C) or the like, for example coupled to the output node of operational amplifier  50 . 
     When system  10  is operating at steady state, the high gain of amplifier  50  and the effect of feedback loop  60  will force the V CLK  and V CLKB  DC input signals to be substantially equal. Advantageously, when V CLK =V CLKB , duty cycle D CLK =D CLKB . But since D CLK =1−D CLKB , it follows that D CLK =50% and D CLKB =50%, which means duty cycle of the waveforms CLK, CLKB is 50%. Thus, the architecture exemplified by system  10  in FIG. 3 can indeed automatically and continuously act to adjust and maintain the duty cycle of an output clock signal at 50%. Typically, the tolerance of the 50% duty cycle maintained by system  10  will be on the order of at least about ±1% and in practice can be ±0.1%. Thus, it may fairly be said that the duty cycle of the CLKOUT signal output by DCA unit  20  is indeed substantially 50%, notwithstanding that the duty cycle of the CLKIN clock signal to DCA unit  20  may only be nominally 50%, e.g., about 33% to about 67%. 
     Modification and variations may be made to the disclosed embodiments without departing from the subject and spirit of the invention as defined by the following claims.