Patent Publication Number: US-2021181322-A1

Title: Distance measuring device and distance measuring method

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a continuation of application Ser. No. 15/705,997 filed Sep. 15, 2017 and is based upon and claims the benefit of priority from the prior Japanese Patent Applications No. 2017-053379, filed on Mar. 17, 2017 and No. 2017-138306, filed on Jul. 14, 2017; the entire contents of which are incorporated herein by reference. 
    
    
     FIELD 
     Embodiments described herein relate generally to a distance measuring device and a distance measuring method. 
     BACKGROUND 
     In recent years, keyless entry for facilitating unlocking and locking of a car has been adopted in many cars. This technique performs unlocking and locking of a door using communication between a key of an automobile and the automobile. Further, in recent years, a smart entry system that makes it possible to perform, with a smart key, unlocking and locking of a door lock and start an engine without touching a key has been also adopted. 
     However, a lot of incidents occur in which an attacker intrudes into communication between a key and an automobile and steals the automobile. As measures against the attack (so-called relay attack), a measure for measuring the distance between the key and the automobile and, when determining that the distance is equal to or larger than a predetermined distance, it is being reviewed to prohibit control of the automobile by communication. 
     As a distance measuring technique, many techniques exist, such as a two-frequency CW (continuous wave) scheme, an FM (frequency modulated) CW scheme, a Doppler scheme, and a phase detection scheme. In general, in distance measurement, a distance from a measuring device to a target object is calculated by providing a transmitter and a receiver in the same housing of the measuring device, hitting a radio wave emitted from the transmitter against the target object, and detecting a reflected wave of the radio wave with the receiver. 
     However, when it is taking into account a relatively small reflection coefficient of the target object, limitation on output power due to the Radio Law, and the like, in the distance measuring technique for measuring a distance using the reflected wave, a measurable distance is relatively small and is insufficient for use in the measures against the relay attack. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram showing a distance measuring system in which a distance measuring device according to a first embodiment of the present invention is adopted; 
         FIG. 2A  is an explanatory diagram for explaining the principle of distance measurement by a phase detection scheme for detecting a phase of a reflected wave and problems of the distance measurement; 
         FIG. 2B  is an explanatory diagram for explaining the principle of the principle of distance measurement by a phase detection scheme for detecting a phase of a reflected wave and the problems of the distance measurement; 
         FIG. 3A  is an explanatory diagram for explaining problems of the distance measurement by the phase detection scheme; 
         FIG. 3B  is an explanatory diagram for explaining the problems of the distance measurement by the phase detection scheme; 
         FIG. 4  is a circuit diagram showing an example of specific configurations of a transmitting section  14  and a receiving section  15  shown in  FIG. 1 ; 
         FIG. 5  is a circuit diagram showing an example of specific configuration of a transmitting section  24  and a receiving section  25  shown in  FIG. 1 ; 
         FIG. 6  is a flowchart for explaining operation in the first embodiment; 
         FIG. 7  is an explanatory diagram for explaining a method of calculating a distance using a system of residue; 
         FIG. 8  is an explanatory diagram for explaining the method of calculating a distance using the system of residue; 
         FIG. 9  is an explanatory diagram showing an example in which a distance is plotted on the horizontal axis and a phase is plotted on the vertical axis, in which three wave signals having different angular frequencies each other are transmitted; 
         FIG. 10  is an explanatory diagram for explaining a method of selecting a correct distance through amplitude observation of a detected signal; 
         FIG. 11A  is a flowchart for explaining a second embodiment; 
         FIG. 11B  is an explanatory diagram for explaining the second embodiment; 
         FIG. 11C  is an explanatory diagram for explaining the second embodiment; 
         FIG. 12A  is an explanatory diagram for explaining the second embodiment; 
         FIG. 12B  is an explanatory diagram for explaining the second embodiment; 
         FIG. 13  is an explanatory diagram for explaining the second embodiment; 
         FIG. 14  is an explanatory diagram for explaining the second embodiment; 
         FIG. 15  is an explanatory diagram for explaining the second embodiment; 
         FIG. 16  is a block diagram showing a third embodiment of the present invention; 
         FIG. 17  is an explanatory diagram showing a fourth embodiment of the present invention; 
         FIG. 18  is an explanatory diagram showing the fourth embodiment of the present invention; 
         FIG. 19  is an explanatory diagram showing one of various sequences; 
         FIG. 20  is an explanatory diagram showing one of the various sequences; 
         FIG. 21  is an explanatory diagram showing one of the various sequences; 
         FIG. 22  is an explanatory diagram showing one of the various sequences; 
         FIG. 23  is an explanatory diagram showing one of the various sequences; 
         FIG. 24  is an explanatory diagram showing one of the various sequences; 
         FIG. 25  is an explanatory diagram showing one of the various sequences; 
         FIG. 26  is an explanatory diagram showing one of the various sequences; 
         FIG. 27  is an explanatory diagram showing one of the various sequences; 
         FIG. 28  is an explanatory diagram showing one of the various sequences; 
         FIG. 29  is an explanatory diagram showing one of the various sequences; 
         FIG. 30  is an explanatory diagram showing one of the various sequences; 
         FIG. 31  is an explanatory diagram showing one of the various sequences; 
         FIG. 32  is an explanatory diagram showing one of the various sequences; 
         FIG. 33  is an explanatory diagram showing one of the various sequences; 
         FIG. 34  is an explanatory diagram showing one of the various sequences; 
         FIG. 35  is an explanatory diagram showing one of the various sequences; 
         FIG. 36  is an explanatory diagram showing one of the various sequences; 
         FIG. 37  is an explanatory diagram showing one of the various sequences; 
         FIG. 38  is an explanatory diagram showing one of the various sequences; 
         FIG. 39  is an explanatory diagram showing one of the various sequences; 
         FIG. 40  is an explanatory diagram showing one of the various sequences; 
         FIG. 41  is an explanatory diagram showing one of the various sequences; 
         FIG. 42  is an explanatory diagram showing one of the various sequences; 
         FIG. 43  is an explanatory diagram showing one of the various sequences; 
         FIG. 44  is an explanatory diagram showing one of the various sequences; 
         FIG. 45  is an explanatory diagram showing a relation between a transmission sequence and a period in which an initial phase is maintained; 
         FIG. 46  is an explanatory diagram showing a carrier frequency used for distance measurement; 
         FIG. 47  is a flowchart for explaining a modification; 
         FIG. 48A  is an explanatory diagram showing, in a simplified manner, an example of the configurations of an oscillator  13 , the transmitting section  14 , and the receiving section  15  of a device  1 ; 
         FIG. 48B  is an explanatory diagram showing, in a simplified manner, an example of the configurations of an oscillator  23 , the transmitting section  24 , and the receiving section  25  of a device  2 ; 
         FIG. 49A  is an explanatory diagram showing, in a simplified manner, an example of the configurations of the oscillator  13 , the transmitting section  14 , and the receiving section  15  of the device  1 ; 
         FIG. 49B  is an explanatory diagram showing, in a simplifier manner, an example of the configurations of the oscillator  23 , the transmitting section  24 , and the receiving section  25  of the device  2 ; 
         FIG. 50  is a circuit diagram more specifically showing an example of a circuit that generates signals given to multipliers TM 11  and TM 12  in  FIG. 4 ; 
         FIG. 51  is a circuit diagram more specifically showing an example of a circuit that generates signals given to multipliers TM 21  and TM 22  in  FIG. 5 ; 
         FIG. 52  is a circuit diagram showing an example of specific configurations of the transmitting section  14  and the receiving section  15  shown in  FIG. 1 ; 
         FIG. 53  is a circuit diagram showing an example of specific configurations of the transmitting section  24  and the receiving section  25  shown in  FIG. 1 ; 
         FIG. 54  is a circuit diagram showing an example of the specific configurations of the transmitting section  14  and the receiving section  15  shown in  FIG. 1 ; 
         FIG. 55  is a circuit diagram showing an example of the specific configurations of the transmitting section  24  and the receiving section  25  shown in  FIG. 1 ; 
         FIG. 56  is a flowchart for explaining an example corresponding to  FIG. 11A  in which a second device transmits phase information to a first device; and 
         FIG. 57  is a flowchart for explaining an example corresponding to  FIG. 47 . 
     
    
    
     DETAILED DESCRIPTION 
     A distance measuring device according to an embodiment is a distance measuring device that calculates a distance on a basis of carrier phase detection, the distance measuring device including a calculating section configured to calculate, on a basis of phase information acquired by a first device and a second device, at least one of which is movable, a distance between the first device and the second device. The first device includes: a first reference signal source; and a first transceiver configured to transmit two or more first carrier signals and receives two or more second carrier signals using an output of the first reference signal source. The second device includes: a second reference signal source configured to operate independently from the first reference signal source; and a second transceiver configured to transmit the two or more second carrier signals and receives the two or more first carrier signals using an output of the second reference signal source. The calculating section calculates the distance on a basis of a phase detection result obtained by reception of the first and second carrier signals. 
     Embodiments of the present invention are explained below in detail with reference to the drawings. 
     First Embodiment 
       FIG. 1  is a block diagram showing a distance measuring system in which a distance measuring device according to a first embodiment of the present invention is adopted. 
     In the present embodiment, an example is explained in which a phase detection scheme for detecting a phase of an unmodulated carrier is adopted and communication-type distance measurement for calculating a distance between respective devices through communication between the respective devices is adopted. In a general phase detection scheme for detecting a phase of a reflected wave, a measurable distance is relatively short as explained above. Therefore, in the present embodiment, the communication-type distance measurement for performing communication between devices is adopted. However, since respective transmitters of the respective devices independently operate from each other, initial phases of transmitted radio waves from the respective transmitters are different from each other. An accurate distance cannot be calculated by the phase detection scheme in the past for calculating a distance according to a phase difference. Therefore, in the present embodiment, as explained below, phase information calculated by reception of one device is transmitted to the other device to make it possible to calculate an accurate distance in the other device. 
     First, the principle of distance measurement by the phase detecting scheme for detecting a phase of a reflected wave and problems of the distance measurement are explained with reference to the explanatory diagrams of  FIGS. 2A and 2B . 
     (Phase Detection Scheme) 
     In the phase detection scheme, for distance measurement, signals having two frequencies deviating from a center angular frequency ω C1  by an angular frequency ±ω B1  are transmitted. In a distance measuring device that measures a distance using a reflected wave, a transmitter and a receiver are provided in the same housing. A transmission signal (a radio wave) emitted from the transmitter is reflected on a target object and a reflected wave of the radio wave is received by the receiver. 
       FIGS. 2A and 2B  show this state. A radio wave emitted from a transmitter T is reflected by a wall W and received by a receiver S. 
     As shown in  FIG. 2A , an angular frequency of a radio wave emitted from the transmitter is represented as ω C1 +ω B1  and an initial phase is represented as θ 1H . In this case, a transmission signal (a transmission wave) tx 1 ( t ) emitted from the transmitter is represented by the following Equation (1): 
         tx 1( t )=cos {(ω C1 +ω B1 ) t+θ   1H }  (1)
 
     The transmission signal reaches a target object (a wall W) apart from the transmitter by a distance R with a delay time τ 1  and is reflected and received by the receiver. Since the speed of the radio wave is equal to the speed of light c(=3×10 8  m/s), τ 1 =(R/c) (seconds). The signal received by the receiver delays by 2τ 1  with respect to the emitted signal. Therefore, a received signal (a received wave) rx 1 ( t ) of the receiver is represented by the following Equations (2) and (3): 
         rx 1( t )=cos {(ω C1 +ω B1 ) t+θ   1H −θ 2×Hτ1 }  (2)
 
       θ 2×Hτ1 =(ω C1 +ω B1 )2 τ1   (3)
 
     That is, the transmission signal is received by the receiver with a phase shift of a multiplication result (θ 2×Hτ1 ) of the delay time and the transmission angular frequency. 
     Similarly, as shown in  FIG. 2B , the transmission signal tx 1 ( t ) and the received signal rx 1 ( t ) in the case in which an angular frequency ω C1 −ω B1  is used are represented by the following Equations (4) to (6) with an initial phase set to θ 1L : 
         tx 1( t )=cos {(ω C1 −ω B1 ) t+θ   1L }  (4)
 
         rx 1( t )=cos {(ω C1 −ω B1 ) t+θ   1L −θ 2×Lτi }  (5)
 
       θ 2×Lτ1 =(ω C1 −ω B1 )2 τ1   (6)
 
     When a phase shift amount that occurs until the transmission signal having the angular frequency ω C1 +ω B1  is received is represented as θ H1 (t) and a phase shift amount that occurs until the transmission signal having the angular frequency ω C1 −ω B1  is received is represented as θ L1 (t), a difference between phase shifts of the two received waves is represented by the following Equation (7) obtained by subtracting Equation (6) from Equation (3): 
       θ H1 ( t )−θ L1 ( t )=(θ 2×Hτ1 −θ 2×Lτ1 )=2ω B1 ×2 τ1   (7)
 
     where τ 1 =R/c. Since the differential frequency ω B1  is known, if the difference between the phase shift amounts of the two received waves is measured, the distance R can be calculated as follows from a measurement result: 
         R=c ×(θ 2×Hτ1 −θ 2×Lτ1 )/(4 ωB1 )
 
     Incidentally, in the above explanation, the distance R is calculated taking into account only the phase information. Amplitude is examined below concerning a case in which a transmission wave having the angular frequency ω C1 +ω B1  is used. The transmission wave indicated by Equation (1) described above delays by a delay amount τ 1 =R/c at a point in time when the transmission wave reaches a target object away from the transmitter by the distance R. Amplitude is attenuated by attenuation L 1  corresponding to the distance R. The transmission wave changes to a wave rx 2 ( t ) represented by the following Equation (8): 
         rx 2( t )= L   1  cos {(ω C1 +ω B1 ) t+θ   1H −(ω C1 +ω B1 )τ 1 }  (8)
 
     Further, the transmission wave is attenuated by attenuation L RFL  when the transmission wave is reflected from the target object. A reflected wave tx 2 ( t ) in the target object is represented by the following Equation (9): 
         tx 2( t )= L   RFL   L   1  cos {(ω C1 +ω B1 ) t+θ   1H −(ω C1 +ω B1 )τ 1 }  (9)
 
     The received signal rx 1 ( t ) received by the receiver is delayed by a delay amount τ 1 =R/c(s) from the target object. Amplitude is attenuated by attenuation L 1  corresponding to the distance R. Therefore, the received signal is represented by the following Equation (10): 
         rx 1( t )= L   1   ×L   RFL   ×L   1  cos {(ω C1 +ω B1 ) t+θ   1H −2(ω C1 +ω B1 )τ 1 }  (10)
 
     In this way, the transmission signal from the transmitter is attenuated by L 1 ×L RFL ×L 1  until the transmission signal reaches the receiver. Signal amplitude that can be emitted from the transmitter in distance measurement needs to conform to the Radio Law according to an applied frequency. For example, a specific frequency in a 920 MHz band involves limitation to suppress transmission signal power to 1 mW or less. From the viewpoint of a signal-to-noise ratio of the received signal, it is necessary to suppress attenuation between transmission and reception in order to accurately measure a distance. However, as explained above, since attenuation is relatively large in the distance measurement for measuring a distance using a reflected wave, a distance that can be accurately measured is short. 
     Therefore, as explained above, in the present embodiment, by transmitting and receiving signals between the two devices without using a reflected wave, attenuation is reduced by L RFL ×L 1  to increase the distance that can be accurately measured. 
     However, the two devices are apart from each other by the distance R and cannot share the same reference signal. In general, it is difficult to synchronize the transmission signal with a local oscillation signal used for reception. That is, between the two devices, deviation occurs in a signal frequency and an initial phase is unknown. Problems in distance measurement performed using such an asynchronous transmission wave are explained. 
     (Problems in the Case of Asynchronization) 
     In the distance measuring system in the present embodiment, in distance measurement between two objects, two devices (a first device and a second device) that emit carrier signals (transmission signals) asynchronously from each other are disposed in the positions of the respective objects and the distance R between the two devices is calculated. In the present embodiment, carrier signals having two frequencies deviating from a center angular frequency ω C1  by the angular frequency ±ω B1  are transmitted in the first device. Carrier signals having two frequencies deviating from the center angular frequency ω C2  by an angular frequency ±ω B2  are transmitted in the second device. 
       FIGS. 3A and 3B  are explanatory diagrams for explaining problems in the case in which the phase detection scheme is simply applied between two devices A 1  and A 2 . It is assumed that a transmission signal of the device A 1  is received by the device A 2 . A local oscillator of the device A 1  generates a signal having a frequency necessary for generating, in a heterodyne scheme, two transmission waves having carrier angular frequencies ω C1 +ω B1  and ω C1 −ω B1 . The device A 1  transmits two transmission waves having the angular frequencies. A local oscillator of the device A 2  generates a signal having a frequency necessary for generating, in a heterodyne scheme, two transmission waves having carrier angular frequencies ω C2 +ω B2  and ω C2 −ω B2 . The device A 2  performs reception in the heterodyne scheme using the signal generated by the local oscillator of the device A 2 . 
     The distance between the transmission device and the reception device is represented as  2 R to correspond to the distance in the case in which the reflected wave is used. Initial phases of a transmission signal having the angular frequency ω C1 +ω B1  and a transmission signal having the angular frequency ω C1 −ω B1  transmitted from the device A 1  are respectively represented as θ 1H  and θ 1L . Initial phases of two signals having the angular frequencies ω C2 +ω B2  and ω C2 −ω B2  of the device A 2  are respectively represented as  02 H and  021 L. 
     First, a phase is considered concerning the transmission signal having the angular frequency ω C1 +ω B1 . The transmission signal represented by Equation (1) described above is output from the device A 1 . The received signal rx 2 ( t ) in the device A 2  is represented by the following Equation (11): 
         rx 2( t )=cos {(ω C1 +ω B1 ) t+θ   1H −θ 2×Hτ1 }  (11)
 
     The device A 2  multiplies together two signals cos {(ω C2 +ω B2 )t+θ 2H } and sin{(ω C2 +ω B2 )t+θ 2H } and a received wave of Equation (11) to thereby separate the received wave into an in-phase component (an I signal) and a quadrature component (a Q signal). A phase of the received wave (hereinafter referred to as detected phase or simply referred to as phase) can be easily calculated from the I and Q signals. That is, a detected phase θ H1 (t) is represented by the following Equation (12). Note that, in the following Equation (12), since a term of harmonics near an angular frequency ω C1 +ω c2  is removed during demodulation, the term is omitted. 
       θ H1 ( t )=tan −1 ( Q ( t )/ I ( t ))=−{(ω C1 −ω C2 ) t +(ω B1 −ω B2 ) t+θ   1H −θ 2×Hτ1 }  (12)
 
     Similarly, when the transmission signal having the angular frequency ω C1 −ω B1  is transmitted from the device A 1 , a detected phase θ L1 (t) calculated from the I and Q signals obtained in the device A 2  is represented by the following Equation (13). Note that, in the following Equation (13), since a term of harmonics near the angular frequency ω C1 +ω C2  is removed during demodulation, the term is omitted. 
       θ L1 ( t )=tan −1 ( Q ( t )/ I ( t ))=−{(ω C1 −ω C2 ) t −(ω B1 −ω B2 ) t+θ   1L −θ 2L −ω 2×Hτi }  (13)
 
     A phase difference between these two detected phases (hereinafter referred to as detected phase difference or simply referred to as phase difference) θ H1 (t)−θ L1 (t) is represented by the following Equation (14): 
       θ H1 ( t )−θ L1 ( t )=−2(ω B1 −ω B2 ) t +(θ 1H −θ 1L )−(θ 2H −θ 2L )+(θ 2×Hτ1 −θ 2×Lτ1 )  (14)
 
     In the distance measuring device in the past that measures a distance using a reflected wave, the device A 1  and the device A 2  are the same device and share the local oscillator. Therefore, the following Equations (15) to (17) are satisfied: 
       ω B1 =ω B2   (15)
 
       θ 1H =θ 2H   (16)=
 
       θ 1L =θ 2L   (17)
 
     When Equations (15) to (17) hold, Equation (14) is equal to Equation (7) described above. The distance R between the device A 1  and the device A 2  can be calculated according to a phase difference calculated by I and Q demodulation processing for the received signal in the device A 2 . 
     However, since the device A 1  and the device A 2  are provided to be separated from each other and the local oscillators operate independently from each other, Equations (15) to (17) described above are not satisfied. In this case, unknown information such as a difference between initial phases is included in Equation (14). A distance cannot be correctly calculated. 
     (Distance Measuring Method of the Embodiment) 
     The signals having the two angular frequencies explained above transmitted by the first device are received in the second device and phases of the respective signals are calculated. The signals having the two angular frequencies explained above transmitted by the second device are received in the first device and phases of the respective signals are calculated. Further, phase information is transmitted from either one of the first device and the second device to the other. In the present embodiment, as explained below, the distance R between the first device and the second device is calculated by adding up a phase difference between the two signals calculated by the reception of the first device and a phase difference between the two signals calculated by the reception of the second device. Note that the phase information may be the I and Q signals or may be information concerning phases calculated from the I and Q signals or may be information concerning a difference between phases calculated from two signals having different frequencies. 
     (Configuration) 
     In  FIG. 1 , the first device  1  (hereinafter referred to as device  1  as well) and the second device  2  (hereinafter referred to as device  2  as well) are disposed to be separated from each other by the distance R. At least one of the device  1  and the device  2  is movable. The distance R changes according to the movement. A control section  11  is provided in the device  1 . The control section  11  controls respective sections of the device  1 . The control section  11  is configured of a processor including a CPU. The control section  11  may operate according to a computer program stored in a not-shown memory and control the respective sections. 
     An oscillator  13  is controlled by the control section  11  and generates oscillation signals (local signals) having two frequencies on a basis of a reference oscillator incorporated in the oscillator  13 . The respective oscillation signals from the oscillator  13  are supplied to a transmitting section  14  and a receiving section  15 . Angular frequencies of the oscillation signals generated by the oscillator  13  are set to angular frequencies necessary for generating two waves of ω C1 +ω B1  and ω C1 −ω B1  as angular frequencies of transmission waves of the transmitting section  14 . 
     The transmitting section  14  can be configured of, for example, a quadrature modulator. The transmitting section  14  is controlled by the control section  11  to be capable of outputting two transmission waves of a transmission signal having the angular frequency ω C1 +ω B1  and a transmission signal having the angular frequency ω C1 −ω B1 . The transmission waves from the transmitting section  14  are supplied to an antenna circuit  17 . 
     The antenna circuit  17  includes one or more antennas and can transmit the transmission waves transmitted from the transmitting section  14 . The antenna circuit  17  receives transmission waves from the device  2  explained below and supplies received signals to the receiving section  15 . 
     The receiving section  15  can be configured of, for example, a quadrature demodulator. The receiving section  15  is controlled by the control section  11  to be capable of receiving and demodulating a transmission wave from the device  2  using, for example, signals having angular frequencies foci and com from the oscillator  13  and separating and outputting an in-phase component (an I signal) and a quadrature component (a Q signal) of the received wave. 
     A configuration of the device  2  is the same as the configuration of the device  1 . That is, a control section  21  is provided in the second device. The control section  21  controls respective sections of the device  2 . The control section  21  is configured of a processor including a CPU. The control section  21  may operate according to a computer program stored in a not-shown memory and control the respective sections. 
     An oscillator  23  is controlled by the control section  21  to generate oscillation signals having two frequencies on a basis of a reference oscillator incorporated in the oscillator  23 . The respective oscillation signals from the oscillator  23  are supplied to a transmitting section  24  and a receiving section  25 . Angular frequencies of the oscillation signals generated by the oscillator  23  are set to angular frequencies necessary for generating two waves of ω C2 +ω B2  and ω C2 −ω B2  as angular frequencies of transmission waves of the transmitting section  24 . 
     The transmitting section  24  can be configured of, for example, a quadrature modulator. The transmitting section  24  is controlled by the control section  21  to be capable of outputting two transmission waves of a transmission signal having an angular frequency ω C2 +ω B2  and a transmission signal having an angular frequency ω C2 −ω B2 . The transmission waves from the transmitting section  24  are supplied to an antenna circuit  27 . 
     The antenna circuit  27  includes one or more antennas and can transmit the transmission waves transmitted from the transmitting section  24 . The antenna circuit  27  receives transmission waves from the device  1  and supplies received signals to the receiving section  25 . 
     The receiving section  25  can be configured of, for example, a quadrature demodulator. The receiving section  25  is controlled by the control section  21  to be capable of receiving and demodulating a transmission wave from the device  1  using, for example, signals having angular frequencies ω C2  and ω B2  from the oscillator  23  and separating and outputting an in-phase component (an I signal) and a quadrature component (a Q signal) of the received wave. 
       FIG. 4  is a circuit diagram showing an example of specific configurations of the transmitting section  14  and the receiving section  15  shown in  FIG. 1 .  FIG. 5  is a circuit diagram showing an example of specific configurations of the transmitting section  24  and the receiving section  25  shown in  FIG. 1 .  FIGS. 4 and 5  show a transceiver of an image suppression scheme. However, the transceiver is not limited to the configuration. 
     Note that a configuration of the image suppression scheme is publicly known. As characteristics of the image suppression scheme, when a higher angular frequency band is demodulated centering on a local angular frequency for a high frequency, that is, ω C1  or ω C2 , a signal in a lower angular frequency band is attenuated and, when a lower angular frequency band is demodulated, a signal in a higher angular frequency band is attenuated. This filtering effect is due to signal processing. The same applies to transmission. When the higher angular frequency band is demodulated centering on ω C1  or ω C2 , sin(ω B1 t+ω B1 ) or sin(ω B2 t+ω B2 ) shown in  FIGS. 4 and 5  is used. When the lower angular frequency band is demodulated, −sin(ω B1 t+ω B1 ) or −sin(ω B2 t+ω B2 ) shown in  FIGS. 4 and 5  is used. The frequency band demodulated is decided by change of such polarity. 
     Note that, in a receiver of the image suppression scheme, a term of harmonics near the angular frequency ω C1 −ω C2  is removed during demodulation. Therefore, in an operation explained below, this term is omitted. 
     The transmitting section  14  is configured of multipliers TM 11  and TM 12  and an adder TS 11 . Oscillation signals having an angular frequency ω C1  and having phases 90 degrees different from each other are respectively given to the multipliers TM 11  and TM 12  from the oscillator  13 . Oscillation signals having an angular frequency ω B1  and having phases 90 degrees different from each other are respectively given to the multipliers TM 11  and TM 12  from the oscillator  13 . An inverted signal of the oscillation signal having the angular frequency ω B1  is also given to the multiplier TM 12  from the oscillator  13 . 
     The multipliers TM 11  and TM 12  respectively multiply together the two inputs and give multiplication results to the adder TS 11 . The adder TS 11  adds up outputs of the multipliers TM 11  and TM 12  and outputs an addition result as a transmission wave tx 1 . 
     The receiving section  15  is configured of multipliers RM 11  to RM 16  and adders RS 11  and RS 12 . A transmission wave of the device  2  is input to the multipliers RM 11  and RM 12  via the antenna circuit  17  as a received signal rx 1 . Oscillation signals having the angular frequency ω C1  and phases 90 degrees different from each other are respectively given to the multipliers RM 11  and RM 12  from the oscillator  13 . The multiplier RM 11  multiplies together the two inputs and gives a multiplication result to the multipliers RM 13  and RM 14 . The multiplier RM 12  multiplies together the two inputs and gives a multiplication result to the multipliers RM 15  and RM 16 . 
     An oscillation signal having the angular frequency (a local angular frequency for baseband processing) ω B1  is given to the multipliers RM 13  and RM 15  from the oscillator  13 . The multiplier RM 13  multiplies together the two inputs and gives a multiplication result to the adder RS 11 . The multiplier RM 14  multiplies together the two inputs and gives a multiplication result to the adder RS 12 . 
     An oscillation signal having the angular frequency ω B1  or an inverted signal of the oscillation signal, that is, a signal orthogonal to the oscillation signal having the angular frequency ω B1  given to the multiplier RM 13  is given to the multipliers RM 14  and RM 16  from the oscillator  13 . The multiplier RM 14  multiplies together the two inputs and gives a multiplication result to the adder RS 12 . The multiplier RM 16  multiplies together the two inputs and gives a multiplication result to the adder RS 11 . 
     The adder RS 11  adds up outputs of the multipliers RM 13  and RM 16  and outputs an addition result as an I signal. The adder RS 12  adds up outputs of the multipliers RM 14  and RM 15  and outputs an addition result as a Q signal. The I and Q signals from the receiving section  15  are supplied to the control section  11 . 
     The circuits shown in  FIGS. 4 and 5  are the same circuit. That is, in  FIG. 5 , the configurations of the multipliers TM 21 , TM 22 , and RM 21  to RM 26  and the adders TS 21 , RS 21 , and RS 22  are respectively the same as the configurations of the multipliers TM 11 , TM 12 , and RM 11  to RM 16  and the adders TS 11 , RS 11 , and RS 12  shown in  FIG. 4 . The configurations are only different in that, since the frequency and the phase of the oscillation signal of the oscillator  23  are different from the frequency and the phase of the oscillation signal of the oscillator  13 , in  FIG. 5 , a local angular frequency for baseband ω B2  is input instead of the angular frequency ω B1  shown in  FIG. 4  and ω C2  is input instead of the angular frequency foci shown in  FIG. 4 . The I and Q signals from the receiving section  25  are supplied to the control section  21 . 
     In the present embodiment, the control section  11  of the device  1  controls the transmitting section  14  to transmit two transmission waves having angular frequencies ω C1 +ω B1  and ω C1 −ω B1  via the antenna circuit  17 . 
     On the other hand, the control section  21  of the device  2  controls the transmitting section  24  to transmit two transmission waves having angular frequencies ω C2 +ω B2  and ω C2 −ω B2  via the antenna circuit  27 . 
     The control section  11  of the device  1  controls the receiving section  15  to receive the two transmission waves from the device  2  and acquires the I and Q signals. The control section  11  calculates a difference between two phases calculated from the I and Q signals respectively obtained by two received signals. 
     Similarly, the control section  21  of the device  2  controls the receiving section  25  to receive the two transmission waves from the device  1  and acquires the I and Q signals. The control section  21  calculates a difference between two phases calculated from the I and Q signals respectively obtained by two received signals. 
     In the present embodiment, the control section  11  of the device  1  gives phase information based on the acquired I and Q signals to the transmitting section  14  and causes the transmitting section  14  to transmit the phase information. Note that, as explained above, as the phase information, for example, a predetermined initial value may be given. The phase information may be I and Q signals calculated from the two received signals, may be information concerning phases calculated from the I and Q signals, or may be information concerning a difference between the phases. 
     For example, the control section  11  may generate I and Q signals based on phase information of a received signal having an angular frequency ω B2  and supplies the I and Q signals respectively to the multipliers TM 11  and TM 12  to transmit the phase information. 
     During output of the oscillation signal having the angular frequency ω B1 , the control section  11  may generate I and Q signals obtained by adding phase information of the received signal having the angular frequency ω B2  to an initial phase of the oscillation signal having angular frequency ω B1  and supply the I and Q signals respectively to the multipliers TM 11  and TM 12  to transmit the phase information. 
     The receiving section  25  of the device  2  receives the phase information transmitted by the transmitting section  14  via the antenna circuit  27 . The receiving section  25  demodulates a received signal and obtains I and Q signals of the phase information. The I and Q signals are supplied to the control section  21 . The control section  21  obtains, according to the phase information from the receiving section  25 , a value including the phase difference acquired by the control section  11  of the device  1 . The control section  21  functioning as a calculating section adds up the phase difference obtained by the reception result of the receiving section  25  and the phase difference based on the phase information transmitted from the device  2  to calculate the distance R between the first device  1  and the second device  2 . 
     Note that, in  FIG. 1 , an example is shown in which both of the first device  1  and the second device  2  have a function of transmitting phase information and a function of giving received phase information to the control section and calculating the distance R. However, it is sufficient that one of the first device  1  and the second device  2  has the function of transmitting phase information and the other has the function of giving received phase information to the control section and calculating the distance R. 
     An operation in the configuration configured as explained above is explained with reference to a flowchart of  FIG. 6 . In  FIG. 6 , an operation of the device  1  is shown on a left side and an operation of the device  2  is shown on a right side. In  FIG. 6 , an arrow connecting steps of the devices  1  and  2  indicates that communication is performed between the devices  1  and  2 . Note that steps S 4 , S 5 , S 14 , and S 15  are substantially simultaneously executed. 
     In step S 1 , the control section  11  of the device  1  determines whether an instruction for a distance measurement start is received. When the instruction for the distance measurement start is received, the control section  11  controls the oscillator  13  to start an output of a necessary oscillation signal. In step S 11 , the control section  21  of the device  2  determines whether an instruction for a distance measurement start is received. When the instruction for the distance measurement start is received, the control section  21  controls the oscillator  23  to start an output of a necessary oscillation signal. 
     Note that, as explained below, in step S 9 , the control section  11  ends oscillation. In step S 20 , the control section  21  ends oscillation. Control of a start and an end of oscillation in the control sections  11  and  21  indicates that oscillation of the oscillators  13  and  23  is not stopped during transmission and reception periods for distance measurement. Actual start and end timings of the oscillation are not limited to the flow shown in  FIG. 6 . In a period in which the oscillation of the oscillators  13  and  23  continues, initial phases of the respective oscillators  13  and  23  are not set anew. 
     The control section  11  of the device  1  generates two transmission signals in step S 3  and causes the antenna circuit  17  to transmit the transmission signals as transmission waves (step S 4 ). The control section  21  of the device  2  generates two transmission signals in step S 13  and causes the antenna circuit  27  to transmit the transmission signals as transmission waves (step S 14 ). 
     It is assumed that an initial phase of an oscillation signal having the frequency ω C1  output from the oscillator  13  of the device  1  is θ c1  and an initial phase of an oscillation signal having the frequency ω B1  is θ B1 . Note that, as explained above, the initial phases θ c1  and θ B1  are not set anew as long as the oscillation of the oscillator  13  continues. 
     Note that it is assumed that an initial phase of an oscillation signal having the frequency ω C2  output from the oscillator  23  of the device  2  is θ c2  and an initial phase of an oscillation signal having the frequency ω B2  is θ B2 . The initial phases θ c2  and θ B2  are not set anew as long as the oscillation of the oscillator  23  continues. 
     Note that, when simultaneous transmission and simultaneous reception of two frequencies are assumed, two wireless sections shown in  FIG. 4  are necessary in the device  1  and two wireless sections shown in  FIG. 5  are necessary in the device  2 . Alternatively, a radio of a superheterodyne scheme or the like is used. However, the respective oscillators use the same radio section. 
     (Transmission and Reception of a Transmission Wave Having the Angular Frequency ω C1 +ω B1  from the Device  1 ) 
     Two transmission waves having the angular frequencies ω C1 +ω B1  and ω C1 −ω B1  are output from the transmitting section  14  of the device  1 , and the transmitting section  14  is composed of the multipliers TM 11  and TM 12  and the adder TS 11 . The transmission signal tx 1 ( t ) having the angular frequency ω C1 +ω B1  is represented by the following Equation (18): 
         tx 1( t )=cos(ω C1   t+θ   C1 )cos(ω B1   t+θ   B1 )−sin(ω C1   t+θ   c1 )sin(ω B1   t+θ   B1 )=cos {(ω C1 +ω B1 ) t+θ   C1 +θ B1 }  (18)
 
     When the distance between the devices  1  and  2  is represented as R and a delay until a transmission wave from the device  1  is received by the device  2  is represented as τ 1 , the received signal rx 2 ( t ) of the device  2  can be represented by the following Equations (19) and (20): 
         rx 2( t )=cos {(ω C1 +ω B1 )( t−τ   1 )+θ C1 +θ B1 }=cos {(ω C1 +ω B1 ) t+θ   C1 +θ B1 −θ τH1 }  (19)
 
       θ τH1 =(ω C1 +ω B1 )τ 1   (20)
 
     The received signal rx 2 ( t ) is received by the antenna circuit  27  and supplied to the receiving section  25 . In the receiver shown in  FIG. 5 , the received signal rx 2 ( t ) is input to the multipliers RM 21  and RM 22 . Subsequently, signals in respective nodes of the receiver shown in  FIG. 5  are sequentially calculated. Outputs of the multipliers RM 21 , RM 23 , and RM 24  are respectively represented as I 1 (t), I 2 (t), and I 3 (t), outputs of the multipliers RM 22 , RM 26 , and RM 25  are respectively represented as Q 1 (t), Q 2 (t), and Q 3 (t), and outputs of the adders RS 21  and RS 22  are respectively represented as I(t) and Q(t). The outputs are represented by the following Equations (21) to (26): 
         I   1 ( t )=cos(ω C2   t−θ   C2 )×cos {(ω C1 +ω B1 ) t+θ   c1 +θ B1 −θ τH1 }  (21)
 
         Q   1 ( t )=sin(ω C2   t+θ   C2 )×cos {(ω C1 +θ B1 ) t+θ   c1 +θ B1 −θ τH1 }  (22)
 
         I   2 ( t )= I   1 ( t )×cos(ω B2   t+θ   B2 )  (23)
 
         Q   2 ( t )= Q   1 ( t )×sin(ω B2   t+θ   B2 )  (24)
 
         I   3 ( t )= I   1 ( t )×sin(ω B2   t+θ   B2 )  (25)
 
         Q   3 ( t )= Q   1 ( t )×cos(ω B2   t+θ   B2 )  (26)
 
     An output I(t) of the adder RS 21  is I(t)=I 2 (t)+Q 2 (t). An output Q(t) of the adder RS 22  is Q(t)=I 3 (t)−Q 3 (t). A phase θ H1 (t) obtained from I(t) and Q(t) is represented by the following Equation (27): 
       θ H1 ( t )=tan −1 ( Q ( t )/ I ( t ))=−{(ω C1 −ω C2 ) t +(ω B1 −ω B2 ) t+θ   C1 −θ C2 +θ B1 −θ B2 −θ τH1 }   (27)
 
     (Transmission and Reception of a Transmission Wave Having the Angular Frequency ω C2 +ω B2  from the Device  2 ) 
     Similarly, when the signal tx 2 ( t ) having the angular frequency (ω C2 +ω B2  transmitted from the device  2  is received by the device  1  after a delay τ 2 , a phase θ H2 (t) obtained from the signals I(t) and Q(t) detected by the device  1  is calculated. 
         tx 2( t )=cos(ω C2   t+θ   C2 )cos(ω B2   t+θ   B2 )−sin(ω C2   t+θ   c2 )sin(ω B2   t+θ   B2 )=cos {(ω C2 +ω B2 ) t+θ   C2 +θ B2 }  (28)
 
         rx 1( t )=cos {(ω C2 +ω B2 )( t−τ   2 )+θ C2 +θ B2 }=cos {(ω C2 +ω B2 ) t+θ   C2 +θ B2 −θ τH2 }  (29)
 
       θ τH2 =(ω C2 +ω B2 )τ 2   (30)
 
     The received signal rx 1 ( t ) is received by the antenna circuit  17  and supplied to the receiving section  15 . In the receiver shown in  FIG. 4 , the received signal rx 1 ( t ) is input to the multipliers RM 11  and RM 12 . Subsequently, signals in respective nodes of the receiver shown in  FIG. 4  are sequentially calculated. Outputs of the multipliers RM 11 , RM 13 , and RM 14  are respectively represented as I 1 (t), I 2 (t), and I 3 (t), outputs of the multipliers RM 12 , RM 16 , and RM 15  are respectively represented as Q 1 (t), Q 2 (t), and Q 3 (t), and outputs of the adders RS 11  and RS 12  are respectively represented as I(t) and Q(t). The outputs are represented by the following Equations (31) to (36): 
         I   1 ( t )=cos(ω C1   t+θ   C1 )×cos {((ω C2 +ω B2 ) t+θ   c2 +θ B2 −θ τ     H2     }   (31)
 
         Q   1 ( t )=sin(ω C1   t+θ   C1 )×cos {(ω C2 +ω B2 ) t+θ   c2 +θ B2 −θ τH2 }  (32)
 
         I   2 ( t )= I   1 ( t )×cos(ω B1   t+θ   B1 )  (33)
 
         Q   2 ( t )= Q   1 ( t )×sin(ω B1   t+θ   B1 )  (34)
 
         I   3 ( t )= I   1 ( t )×sin(ω B1   t+θ   B1 )  (35)
 
         Q   3 ( t )= Q   1 ( t )×cos(ω B1   t+θ   B1 )  (36)
 
     An output I(t) of the adder RS 11  is I(t)=I 2 (t)+Q 2 (t). An output Q(t) of the adder RS 12  is Q(t)=I 3 (t)−Q 3 (t). A phase θ H2 (t)=tan −1 (Q(t)/I(t)) obtained from I(t) and Q(t) is represented by the following Equation (37): 
       θ H2 ( t )=(ω C1 −ω C2 ) t +(ω B1 −ω B2 ) t+θ   C1 −θ C2 +θ B1 −θ B2 +θ τH2   (37)
 
     (Transmission and Reception of a Transmission Wave Having the Angular Frequency ω C1 −ω B1  from the Device  1 ) 
     The signal tx 1 ( t ) having the angular frequency ω C1 −ω B1  transmitted from the device  1  is calculated in the same manner. 
         tx 1( t )=cos(ω C1   t+θ   C1 )cos(ω B1   t+θ   B1 )+sin(ω C1   t+θ   c1 )sin(ω B1   t+θ   B1 )=cos {(ω C1 −ω B1 ) t+θ   C1 −θ B1 }  (38)
 
     Since the distance between the devices  1  and  2  is R and the delay time is the received signal rx 2 ( t ) in the device  2  is represented by the following Equations (39) and (40): 
         rx 2( t )=cos {(ω C1 −ω B1 )( t−τ   1 )+θ C1 +θ B1 }=cos {(ω C1 −ω B1 ) t+θ   C1 −θ B1 −θ τL1 }  (39)
 
       θ τL1 =(ω C1 −ω B1 )τ 1   (40)
 
     Signals of the respective nodes of the device  2  can be represented by the following Equations (41) to (47): 
         I   1 ( t )=cos(ω C2   t+θ   C2 )×cos {(ω C1 −ω B1 ) t+θ   c1 −θ B1 −θ τL1 }  (41)
 
         Q   1 ( t )=sin(ω C2   t+θ   C2 )×cos {(ω C1 −ω B1 ) t+θ   c1 −θ B1 θ τL1 }  (42)
 
         I   2 ( t )= I   1 ( t )×cos(ω B2   t+θ   B2 )  (43)
 
         Q   2 ( t )= Q   1 ( t )×−sin(ω B2   t+θ   B2 )  (44)
 
         I   3 ( t )= I   1 ( t )×−sin(ω B2   t+θ   B2 )  (45)
 
         Q   3 ( t )= Q   1 ( t )×cos(ω B2   t+θ   B2 )  (46)
 
     A phase θ H1 (t)=tan −1 (Q(t)/I(t)) detected by the device  2  from I(t)=I 2 (t)−Q 2 (t) obtained from the adder RS 21  and Q(t)=I 3 (t)+Q 3 (t) obtained from the adder RS 22  is represented by the following Equation (47): 
       θ L1 ( t )=tan −1 {( Q ( t )/ I ( t ))=−(ω C1 −ω C2 ) t −(ω B1 −ω B2 ) t+θ   C1 −θ C2 −(θ B1 −θ B2 )−θ τL1 }   (47)
 
     (Transmission and Reception of a Transmission Wave Having the Angular Frequency ω C2 −ω B2  from the Device  2 ) 
     Similarly, when the signal tx 2 ( t ) having the angular frequency ω C2 −ω B2  transmitted from the device  2  is received by the device  1  after a delay τ 2 , a phase θ L2 (t) obtained from I(t) and Q(t) detected by the device  1  is calculated. 
         tx 2( t )=cos(ω C2   t+θ   C2 )cos(ω B2   t+θ   B2 )+sin(ω C2   t+θ   C2 )sin(ω B2   t+θ   B2 )=cos {(ω C2 ω B2 ) t+θ   C2 −θ B2 }  (48)
 
         rx 1( t )=cos {(ω C2 −ω B2 )( t−τ   2 )+θ C2 −θ B2 }=cos {(ω C2 −ω B2 ) t+θ   C2 −θ B2 − τL2 }  (49)
 
       θ τL2 =(ω C2 −ω B2 )τ 2   (50)
 
     Signals of the respective nodes of the device  1  can be represented by the following Equations (53) to (57): 
         I   1 ( t )=cos(ω C1   t+θ   C1 )×cos {(ω C2 −ω B2 ) t+θ   c2 −θ B2 −θ τL2 }  (51)
 
         Q   1 ( t )=sin(ω C1   t+θ   C1 )×cos {(ω C2 −ω B2 ) t+θ   c2 −θ B2 θ τL2 }  (52)
 
         I   2 ( t )= I   1 ( t )×cos(ω B1   t+θ   B1 )  (53)
 
         Q   2 ( t )= Q   1 ( t )×−sin(ω B1   t+θ   B1 )  (54)
 
         I   3 ( t )= I   1 ( t )×−sin(ω B1   t+θ   B1 )  (55)
 
         Q   3 ( t )= Q   1 ( t )×cos(ω B1   t+θ   B1 )  (56)
 
     A phase θ H1 (t)=tan −1 (Q(t)/I(t)) detected by the device  1  from I(t)=I 2 (t)−Q 2 (t) obtained from the adder RS 11  and Q(t)=I 3 (t)+Q 3 (t) obtained from the adder RS 12  is represented by the following Equation (57): 
       θ L2 ( t )=(ω C1 −ω C2 ) t −(ω B1 −ω B2 ) t+θ   C1 −θ C2 −(θ B1 −θ B2 )+θ τL2   (57)
 
     In step S 6  in  FIG. 6 , the control section  11  of the device  1  acquires the I and Q signals received by the receiving section  15 . In step S 7 , the control section  11  calculates the phases θ τH1 (t) and θ τL1 (t) represented by Equations (27) and (47) described above. In step S 16  in  FIG. 6 , the control section  21  of the device  2  acquires the I and Q signals received by the receiving section  25 . In step S 17 , the control section  21  calculates the phases θ τH2 (t) and θ τL2 (t) represented by Equations (37) and (57) described above. 
     The control section  11  gives acquired phase information to the transmitting section  14  and causes the transmitting section  14  to transmit the phase information (step S 8 ). For example, the control section  11  supplies the I and Q signals based on the phase information instead of the oscillation signals supplied to the multipliers TM 11  and TM 12  shown in  FIG. 4 . As described later, the phase information are given to I T , Q T  signals in  FIG. 50  and  FIG. 51 . Note that another transmitter for transmitting the phase information may be used. 
     In step S 18 , the control section  21  of the device  2  receives the phase information from the device  1 . As explained above, the phase information may be the I and Q signals from the receiving section  15  of the device  1 , may be information concerning phases obtained from the I and Q signals, or may be information concerning a difference between the phases. 
     In step S 19 , the control section  21  performs an operation of the following Equation (58) to calculate a distance. The following Equation (58) is an Equation for adding up a difference between Equation (27) and Equation (47) and a difference between Equation (37) and Equation (57). 
       {θ H1 ( t )−θ L1 ( t )}+{θ H2 ( t )−θ L2 ( t )}=(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )  (58)
 
     The following Equations (59) and (60) hold: 
       θ τH1 −θ τL1 =(ω C1 +ω B1 )τ 1 −(ω C1 −ω B1 )τ 1 =2ω B1τ1   (59)
 
       θ τH2 −θ τL2 =(ω C2 +ω B2 )τ 2 −(ω C2 −ω B2 )τ 2 =2ω B2τ2   (60)
 
     The delays τ 1  and τ 2  of radio waves between the devices  1  and  2  are the same irrespective of a traveling direction. Therefore, the following Equation (61) is obtained from Equation (58): 
       {θ H1 ( t )−θ L1 ( t )}+{θ H2 ( t )−θ L2 ( t )}=(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )=2×(ω B1 +ω B2 )τ 1   (61)
 
     Equation (61) described above indicates that a value proportional to a double of the distance R is calculated by addition of a phase difference between two frequencies by the I and Q signals detected by the device  2  and a phase difference between two frequencies by the I and Q signals detected by the device  1 . In general, the angular frequency ω B1  by the oscillator  13  of the device  1  and the angular frequency ω B2  by the oscillator  13  of the device  2  can be matched with an error in the order of several ten ppm. Therefore, the distance R by Equation (61) described above can be calculated at resolution of equal to or higher than at least approximately 1 m. 
     In step S 9 , the control section  11  stops the oscillator  13 . In step S 20 , the control section  21  stops the oscillator  23 . Note that, as explained above, the control sections  11  and  21  only have to continue the oscillation in a period of transmission and reception in steps S 4 , S 5 , S 14 , and S 15 . Start and end timings of the oscillation of the oscillators  13  and  23  are not limited to the example shown in  FIG. 6 . 
     (Calculation of a Distance by a Residue of 2π) 
     Incidentally, when the addition of the phase differences detected by the device  1  and the device  2  is performed, a result of the addition is sometimes equal to or smaller than −π(rad) or larger than π(rad). In this case, it is possible to calculate a correct distance R with respect to a detected phase by calculating a residue of 2π. 
       FIGS. 7 and 8  are explanatory diagrams for explaining a method of calculating a distance using a system of residue. 
     For example, when R=11 m and ω B1 =ω B2 =2π×5 M, a detected phase difference Δθ 12  obtained by the device  1  and a detected phase difference Δθ 21  obtained by the device  2  are respectively as represented by the following Equations (62) and (63): 
       Δθ 12 =θ τH1 −θ τL1 =−1.8849  (62)
 
       Δθ 21 =θ τH2 −θ τL2 =−6.0737  (63)
 
     The following Equation (61a) is obtained from Equation (61) described above: 
       (½)[{Δθ 12 }+Δ{θ 21 }]=(ω B1 +ω B2 )( R/c )  (61a)
 
       FIG. 7  shows a phase relation between Equations (62) and (63) described above. A phase of a sum of Δθ 21  indicated by an arrow on inner most side and Δθ 12  indicated by a second arrow from the inner side rotating in a clockwise direction on a basis of a phase 0 degree is a phase indicated by a third arrow from the inner side. A half angle of this phase is a phase of a thick line indicated by an arrow on the outermost side. 
     From Equation (61a), −0.3993=(ω B1 +ω B2 )(R/c) is obtained. When this Equation is solved, R=−19 m. It is seen that a distance cannot be correctly calculated because a detected phase difference is larger than −π(rad). 
     Therefore, in the present embodiment, in such a case, as shown in  FIG. 8 , 2π is added to both of Δθ 12  and Δθ 21 . That is, a phase of a sum of 2π+Δθ 21  indicated by an arrow on inner most side and 2π+Δθ 12  indicated by a second arrow from the inner side rotating in a counterclockwise direction on a basis of the phase 0 degree is a phase indicated by a third arrow from the inner side. A half angle of this phase is a phase of a thick line indicated by an arrow on the outermost side. 
       2π+(Δθ 12 +Δθ 21 )/2=2.3008
 
     From Equation (61a), R is calculated as R=11 m. 
     Consequently, in the present embodiment, when the detected phase differences are added up, a residue of 2π only has to be calculated to calculate the distance R. Note that the method of using the residue of 2π in the phase addition is applicable in other embodiments. 
     (Selection from a Plurality of Distance Candidates) 
     Incidentally, a detected phase difference exceeding 2π cannot be detected. Therefore, a plurality of distance candidates are present with respect to a calculated detected phase difference. As a method of selecting a correct distance from the plurality of distance candidates, a method of transmitting three transmission waves having different angular frequencies and a method of determining a distance according to received power exist. 
       FIG. 9  is an explanatory diagram showing an example in which a distance is plotted on the horizontal axis and a phase is plotted on the vertical axis and the third transmission wave having a different angular frequency is transmitted. 
     The following Equation (64) is obtained from Equation (61) described above: 
       (½)×{(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )}=(ω B1 +θ B2 )×( R/c )  (64)
 
     When a left side is described as θ det , a relation between the distance R and θ det  is as indicated by a solid line in  FIG. 9 . However, although a sum θ det  of detected phase differences calculated by Equation (64) described above can take a value other than a value between −π(rad) and π(rad), the sum θ det  of the phase differences is a value converted into a value between −π(rad) and π(rad). In general, this is because a phase angle is displayed within a range [−π(rad), π(rad)]. 
     Referring to  FIG. 9 , candidates of a distance by the sum θ det  of the detected phase differences include R 1 , R 2 , and R 3 . The sum θ det  of the detected phase differences is an addition and subtraction result of phases obtained by transmission and reception of respective transmission waves having angular frequencies ω C1 +ω B1 , ω C1 −ω B1 , ω C2 +ω B2 , and ω C2 −ωθ B2 . However, an addition and subtraction result of phases obtained by transmission and reception of transmission waves having angular frequencies ω C1 +ω B1 /Q and ω C2 +ω B2 /Q is considered anew. Q is a rational number satisfying the following Inequality (65): 
         Q&gt; 1  (65)
 
     A relation between detected phases at the new angular frequencies and the distance R can be indicated by a broken line shown in  FIG. 9 . To select a correct distance from the candidates R 1 , R 2 , and R 3  of the distance, a result of the detected phases obtained at the new angular frequencies is referred to. That is, if θ det 1 is detected, the correct distance is determined as the distance R 1 . If θ det 2 is detected, the correct distance is determined as the distance R 2 . Note that, if a coverage of a radio wave is kept small, the inspection by the phase aliasing is unnecessary. Note that the transmission at the different three frequencies is explained above. However, the same can be realized by transmitting different three or more frequencies. 
     A method of selecting a correct distance according to amplitude observation of a detected signal is explained with reference to the explanatory diagram of  FIG. 10 . 
     In Equation (8) described above, the amplitude is attenuated at the attenuation L 1  according to the distance R. However, propagation attenuation in a free space is represented by the following Equation (66): 
         L   1 =(λ/4π R ) 2   (66)
 
     where λ is a wavelength. According to Equation (66), if the distance R is large, the attenuation L 1  is also large and, if the distance R is small, the attenuation L 1  is also small.  FIG. 10  shows this relation. When it is assumed that an antenna gain of transmission and reception is 1 and transmission power is P 0 , received power P 1  at the distance R 1  and received power P 2  at the distance R 2  are respectively represented by the following Equations (67) and (68): 
         P   1 =(λ/4π R   1 ) 2   ×P   0   (67)
 
         P   2 =(λ/4π R   2 ) 2   ×P   0   (68)
 
     It is possible to distinguish the distances R 1  and R 2  from the sum θ det  of the detected phase differences and the received power. 
     Note that, in this case, it is possible to perform sure distance measurement by using the residue of 2π in the phase addition as well. 
     As explained above, in the present embodiment, the signals having the two angular frequencies are transmitted from the first device and the second device to the second device and the first device respectively, and the two phases of the two received signals having different angular frequencies are respectively calculated in the first and second devices. Any one of the first device and the second device transmits calculated phase information to the other. The device that receives the phase information accurately calculates the distance between the first device and the second device irrespective of initial phases of the oscillators of the first device and the second device according to an addition result of a phase difference between the two received signals received by the first device and a phase difference between the two received signals received by the second device. In the distance measuring system, a reflected wave is not used. The accurate distance measurement is performed by only one direction from the first device and the second device. It is possible to increase a measurable distance. 
     Second Embodiment 
       FIG. 11A  is a flowchart according to a second embodiment of the present invention.  FIGS. 11B to 15  are explanatory diagrams according to the second embodiment. A hardware configuration in the present embodiment is the same as the hardware configuration in the first embodiment. A method of transmitting two transmission waves and receiving the transmission waves in both the devices  1  and  2  and calculating a distance according to an addition result of phase differences detected from received signals is the same as the method in the first embodiment. 
       {θ H1 ( t )−θ L1 ( t )}+{θ H2 ( t )−θ L2 ( t )}=(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )  (58)
 
     In the first embodiment, Equation (61) described above for calculating a distance from addition of detected phase differences is calculated assuming that the delays τ 1  and τ 2  of the radio wave are the same in Equation (58) described above. However, Equation (58) is an example in the case in which transmission and reception processing is simultaneously performed in the devices  1  and  2 . 
     However, because of the provision of the Radio Law in the country, a frequency band in which simultaneous transmission and reception cannot be performed is present. For example, a 920 MHz band is an example of the frequency band. When distance measurement is performed in such a frequency band, transmission and reception has to be performed in time-series processing. In the present embodiment, an example adapted to such time-series transmission and reception is explained. 
     (Problems in the Time-Series Transmission and Reception) 
     When it is specified that only one wave can be transmitted and received at the same time between the devices  1  and  2 , it is necessary to carry out, in time-series processing, transmission and reception of at least four waves necessary for distance measurement. However, when the time-series transmission and reception is carried out, a phase equivalent to a delay that occurs in time-series processing is added to a detected phase. A phase required for propagation cannot be calculated. A reason for this is explained by modifying Equation (58) explained above. 
     Note that a broken line portion of  FIG. 6  is substantially simultaneously executed. However, when transmission and reception of one wave is performed in time-series processing, the broken line portion is as shown in  FIG. 11A . 
     As in the first embodiment, in the devices  1  and  2  separated from each other by the distance R, a phase (shift amount) at the time when a signal having the angular frequency ω C1 +ω B1  transmitted from the device  1  is detected in the device  2  is represented as θ H1 , a phase at the time when a signal having the angular frequency ω C1 −ω B1  transmitted from the device  1  is detected in the device  2  is represented as θ L1 , a phase (shift amount) at the time when a signal having the angular frequency ω C2 +ω B2  transmitted from the device  2  is detected in the device  1  is represented as θ H2 , and a phase at the time when a signal having the angular frequency ω C2 −ω B2  transmitted from the device  2  is detected in the device  1  is represented as θ L2 . 
     For example, phase detection order is set as θ H1 , θ L2 , θ H2 , and θ L1 . As shown in  FIGS. 11B and 11C , respective transmission signals are transmitted and received while being shifted by a time T. In this case, a time is substituted in (t) of Equations (27), (37), (47), and (57) described above. The following Equation (120) obtained by modifying Equation (58) described above holds: 
       {θ H1 ( t )−θ L1 ( t+ 3 T )}+{θ H2 ( t+ 2 T )−θ L2 ( t+T )}=(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )+(ω C1 −ω C2 )4 T   (120)
 
     A last term of Equation (120) described above is a phase added by the time-series transmission and reception. The added phase is a multiplication result of error angular frequencies between the local angular frequencies used in the device  1  and the device  2  and a delay 4T, where the local frequencies are almost the same frequencies as the RF frequencies used in the device  1  and the device  2 . When a local frequency is set to 920 MHz, a frequency error is set to 40 ppm, and a delay T is set to 0.1 ms, the added phase is 360°×14.7. It is seen that an error due to the added phase is too large and distance measurement cannot be correctly performed. 
     The phase detection order is set as θ H1 , θ L1 , θ H2 , and θ L2 .  FIGS. 12A and 12B  show an example of this case. In this case, the following Equation (121) is obtained by modifying Equation (58) described above: 
       {θ H1 ( t )−θ L1 ( t+T )}+{θ H2 ( t+ 2 T )−θ L2 ( t+ 3 T )}=(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )+(ω B1 −ω B2 )4 T   (121)
 
     A last term of Equation (121) described above is a phase added by the time-series transmission and reception. The added phase is a multiplication result of error angular frequencies between the local angular frequencies for baseband processing used in the device  1  and the device  2  and a delay 4T, where the local frequencies for baseband processing are almost the same frequencies as the baseband frequencies used in the device  1  and the device  2 . When a local frequency for baseband processing is set to 5 MHz, a frequency error is set to 40 ppm, and the delay T is set to 0.1 ms, the added phase is 360°×0.08=28.8°. It is seen from precedence that distance measurement can be correctly performed. 
     However, in this case, it depends on a system whether an error is within an allowable error of system specifications. The present embodiment presents a time-series procedure for reducing a distance error that occurs because of the time-series transmission and reception. Note that the present embodiment indicates a procedure that takes into account the regulation of transmission and reception specified by the Radio Law. 
     (Specific Procedure) 
     First, an influence due to a transmission delay is considered. 
     The following Equation (122) is obtained by modifying Equation (58) described above: 
       {θ H1 ( t )+θ H2 ( t )}−{θ L1 ( t )+θ L2 ( t )}=(θ τH1 +θ τH2 )−(θ τL1 +θ τL2 )  (122)
 
     In the equation, 
       θ H1 ( t )+θ H2 ( t )=θ τH1 −θ τH2   (123)
 
       θ L1 ( t )+θ L2 ( t )=θ τL1 +θ τL2   (124)
 
     In wireless communication, there is a provision that, when a signal addressed to oneself is received, a reply can be transmitted without carrier sense. According to the provision, after transmission of a signal from the device  1  to the device  2  ends, a reply is immediately transmitted from the device  2  to the device  1 . To simplify an analysis, it is assumed that the device  2  transmits a reply to the device  1  after to from the transmission by the device  1 . The following Equation (125) is obtained from Equations (27) and (37): 
       θ H1 ( t )+θ H2 ( t+t   0 )=θ τH1 +θ τH2 +{(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (125)
 
     A delay t 0  is a shortest time period and includes a time period in which a signal having the angular frequency ω C1 +ω B1  is transmitted from the device  1  to the device  2 , a transmission and reception timing margin, and a propagation delay. A third term and a fourth term on a right side are phase errors due to the delay t 0 . The fourth term is particularly a problem because a frequency is high. This is referred to below. 
     The delay T is further added to a left side of Equation (125).  FIG. 13  shows such a transmission procedure. As shown in  FIG. 13 , an addition value of a detected phase in this case is the same irrespective of the addition of the delay T. Therefore, the following Equation (126) is obtained: 
       θ H1 ( t+T )+θ H2 ( t+t   0   +T )=θ τH1 +θ τH2 +{(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (126)
 
     A right side of Equation (126) described above and a right side of Equation (125) described above are the same. That is, if a relative time difference is the same (in the example explained above, T), an addition result of a phase in which a signal transmitted from the device  1  is received by the device  2  and a phase in which a signal transmitted from the device  2  is received by the device  1  does not change irrespective of the delay T. That is, the addition result of the phases is a value that does not depend on the delay T. 
     Transmission and reception of the angular frequency foci-corn signal between the device  1  and the device  2  is explained the same. That is, the following Equations (127) and (128) are obtained from Equations (47) and (57) described above: 
       θ L1 ( t )+θ L2 ( t+t   0 )=θ τL1 +θ τL2 +{−(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (127)
 
       θ L1 ( t+T )+θ L2 ( t+t   0   +T )=θ τL1 +θ τL2 +{−(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (128)
 
     From the above examination, a sequence is considered in which, after transmission and reception in both directions of the angular frequency ω C1 +ω B1  signal, transmission of reception of the angular frequency ω C1 −ω B1  signal is performed. When a transmission start time of the angular frequency ω C1 −ω B1  signal from the device  1  is represented as T on a basis of a transmission start time of the angular frequency ω C1 +ω B1  signal, the following Equation (129) is obtained from Equations (125) and (128) describe above, where T&gt;t 0  is assumed. 
       θ H1 ( t )+θ H2 ( t+t   0 )−{θ L1 ( t+T )+θ L2 ( t+t   0   +T )}=θ τH1 −θ τL1 +θ τH2 −θ τL2 +2(ω B1 −ω B2 ) t   0   (129)
 
     A last term of a left side of Equation (129) described above is a phase error due to a transmission delay. A delay error due to a received local frequency for high-frequency is cancelled by calculating a difference between the angular frequency ω C1 +ω B1  signal and the angular frequency ω C1 −ω B1  signal. Therefore, the phase error is, in terms of time series, multiplication of a shortest delay time t 0  and an error of a local angular frequency (e.g., 2π×5 MHz) for a baseband processing. If the delay time t 0  is set small, the error is small. Therefore, depending on a value of the delay time t 0 , practically, it is considered possible to perform distance measurement without a problem in accuracy. 
     A method of removing the last term of Equation (129) described above, which is a distance estimation error factor, is explained. 
     The following Equation (130) is obtained from Equations (27) and (37) described above: 
       θ H1 ( t+t   0 )+θ H2 ( t )=θ τH1 +θ τH2 −{(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (130)
 
     Even if a predetermined delay D is added to a left side of Equation (130), as explained above, a value of a right side does not change. Therefore, the following Equation (131) is obtained: 
       θ H1 ( t+t   0   +D )+θ H2 ( t+D )=θ τH1 −θ τH2 −{(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (131)
 
     When the Equations (125) and (131) are added up, the following Equation (132) is obtained: 
       θ H1 ( t )+θ H2 ( t+t   0 )+θ H1 ( t+t   0   +D )+θ H2 ( t+D )=2(θ τH1 +θ τH2 )  (132)
 
     A left side of  FIG. 14  shows a state of Equation (132) described above. When D=t 0  in Equation (132), the following Equation (133) is obtained: 
       θ H1 ( t )+2θ H2 ( t+t   0 )+θ H1 ( t+ 2 t   0 )=2(θ τH1 +θ τH2 )  (133)
 
     A right side of Equation (133) described above is only a term of a radio wave propagation delay corresponding to a distance that does not depend on time. 
     From Equations (47) and (57) described above, the following Equation (134) is obtained: 
       θ L1 ( t+t   0 )+θ L2 ( t )=θ τL1 +θ τL2 −ω B1 −ω B2 )+(ω C1 −ω C2 ) t   0   (134)
 
     Even if the predetermined delay D is added to a left side of Equation (134), a value of a right side does not change. Therefore, the following Equation (135) is obtained: 
       θ L1 ( t+t   0   +D )+θ L2 ( t+D )=θ τL1 +θ τL2 −{−(ω B1 −ω B2 )+(ω C1 −ω C2 )} t   0   (135)
 
     When Equations (127) and (135) described above are added up, the following Equation (136) is obtained: 
       θ L1 ( t )+θ L2 ( t+t   0 )+θ L1 ( t+t   0   +D )+θ L2 ( t+D )=2(θ τL1 +θ τL2 )  (136)
 
     In Equation (136), when D=t 0 , the following Equation (137) is obtained: 
       θ L1 ( t )+2θ L2 ( t+t   0 )+θ L1 ( t+ 2 t   0 )=2(θ τL1 +θ τL2 )  (137)
 
     A right side of Equation (137) described above is only a term of a radio wave propagation delay corresponding to a distance that does not depend on time. 
     Equations (133) and (137) described above mean a sequence for performing phase detection of a transmission signal of the device  1  in the device  2 , performing phase detection of a transmission signal of the device  2  in the device  1  after t 0 , and performing the phase detection of the transmission signal of the device  1  in the device  2  again after 2t 0 . In the following explanation, the process in which transmission of the transmission signal of the device  1  and phase detection in the device  2  for the transmission signal and transmission of the transmission signal of the device  2  and phase detection in the device  1  for the transmission signal alternate and the phase detections are measured again by shifting time is referred to as “repeated alternation”. 
     That is, the repeated alternation for respectively transmitting and receiving two carrier signals in the devices  1  and  2  and transmitting and receiving the carrier signals again at a to interval from the device  1  or  2  to the other device is performed. Consequently, although the order and time of the transmission are limited, it is possible to perform accurate distance measurement that does not depend on time. 
     Further, depending on a transmission and reception sequence of carrier signals, even if the repeated alternation is not performed at the to interval, it is possible to perform accurate distance measurement that does not depend on time. 
     That is, even if a fixed delay T is added to a left side of Equation (136) described above, a right side is fixed. Therefore, the following Equation (138) is obtained: 
       θ L1 ( t+T )+θ 12 ( t+t   0   +T )+θ L1 ( t+t   0   +D+T )+θ L2 ( t+D+T )=2(θ τL1 +θ τL2 )  (138)
 
     The following Equation (139) is obtained from Equations (132) and (138) described above: 
       θ H1 ( t )+θ H2 ( t+t   0 )+θ H1 ( t+t   0   +D )+θ H2 ( t+D )−{θ L1 ( t+T )+θ L2 ( t+t   0   +T )+θ L1 ( t+t   0   +D+T )+θ L2 ( t+D+T )}=2{(θ τH1 −θ τL1 )+(θ τH2 −θ τL2 )}=4×(ω B1 +ω B2 )τ 1   (139)
 
     Equation (139) described above indicates a sequence for, after performing the repeated alternation of reciprocation of the angular frequencies ω C1 +ω B1  signal and ω C2 +ω B2  signal at the time interval D, performing the repeated alternation of reciprocation of the angular frequencies ω C1 −ω B1  signal and ω C2 −ω B2  signal at the time interval D after T from a measurement start. By adopting this sequence, it is possible to remove a distance estimation error factor of the last term of Equation (129) described above and perform accurate distance measurement. 
       FIGS. 14 and 15  show the sequence. It is possible to extract only a propagation delay component by measuring a phase in such a sequence. That is, the control section  11  of the device  1  transmits a transmission wave having the angular frequency ω C1 +ω B1  (hereinafter referred to as transmission wave H 1 A) at predetermined timing. Immediately after receiving the transmission wave H 1 A, the control section  21  of the device  2  transmits a transmission wave having the angular frequency ω C2 +ω B2  (hereinafter referred to as transmission wave H 2 A). Further, after transmitting the transmission wave H 2 A, the control section  21  of the device  2  transmits a transmission wave having the angular frequency ω C2 +ω B2  (hereinafter referred to as transmission wave H 2 B). After receiving the second transmission wave H 2 B, the control section  11  of the device  1  transmits a transmission wave having the angular frequency ω C1 +ω B1  (hereinafter referred to as transmission wave H 1 B). 
     Further, the control section  11  transmits a transmission wave having the angular frequency ω C1 −ω B1  (hereinafter referred to as transmission wave L 1 A). Immediately after receiving the transmission wave L 1 A, the control section  21  of the device  2  transmits a transmission wave having the angular frequency (ω C2 −ω B2  (hereinafter referred to as transmission wave L 2 A). Further, after transmitting the transmission wave L 2 A, the control section  21  of the device  2  transmits a transmission wave having the angular frequency ω C2 −ω B2  (hereinafter referred to as transmission wave L 2 B). After receiving the second transmission wave L 2 B, the control section  11  of the device  1  transmits a transmission wave having the angular frequency ω C1 −ω B1  (hereinafter referred to as transmission wave L 1 B). 
     In this way, as shown in  FIGS. 14 and 15 , the control section  21  of the device  2  acquires a phase θ H1 (t) based on the transmission wave H 1 A in a predetermined time from a predetermined reference time  0 , acquires a phase θ H1 (t+t 0 +D) based on the transmission wave H 1 B in a predetermined time from a time t 0 +D, acquires a phase θ L1 (t+T) based on the transmission wave L 1 A in a predetermined time from the time T, and acquires a phase θ L1 (t+t 0 +D+T) based on the transmission wave L 1 B in a predetermined time from a time t 0 +D+T. 
     The control section  11  of the device  1  acquires a phase θ H2 (t+t 0 ) based on the transmission wave H 2 A in a predetermined time from a time t 0 , acquires a phase θ H2 (t+D) based on the transmission wave H 2 B in a predetermined time from a time D, acquires a phase θ L2 (t+t 0 +T) based on the transmission wave L 2 A in a predetermined time from a time t 0 +T, and acquires a phase θ L2 (t+D+T) based on the transmission wave L 2 B in a predetermined time from a time D+T. 
     At least one of the devices  1  and  2  transmits phase information, that is, calculated four phases or two phase differences or an operation result of Equation (139) described above of the phase differences. The control section of the device  1  or  2 , which receives the phase information, calculates a distance according to an operation of Equation (139) described above. Note that, although “calculate a phase difference” is described in steps S 7  and S 17  in  FIG. 6 , in this case, it is not always necessary to calculate a phase difference in steps S 7  and S 17 . A phase difference may be calculated during the distance calculation in S 19 . 
     In this way, in the present embodiment, by repeatedly alternating the carrier signals from the first device and the second device, even when the carrier signals cannot be simultaneously transmitted and received, it is possible to perform accurate distance measurement. For example, the first device and the second device respectively transmit signals having two angular frequencies twice to the second device and the first device in a predetermined sequence and calculate phase differences respectively in the first and second devices. Any one of the first device and the second device transmits calculated phase information to the other. The device, which receives the phase information, calculates a distance between the first device and the second device on a basis of eight phases calculated by the first device and the second device. Consequently, the distance between the first device and the second device is accurately calculated irrespective of initial phases of the oscillators of the first device and the second device. In this way, even when signals having respective angular frequencies are not simultaneously transmitted and are transmitted and received at timings shifted from each other, it is possible to remove an error of distance estimation and perform accurate distance measurement. 
     Third Embodiment 
       FIG. 16  is a block diagram showing a third embodiment of the present invention.  FIG. 16  shows a transceiver adopted instead of the transceiver by the transmitting section  14  and the receiving section  15  shown in  FIG. 1  and the transceiver by the transmitting section  24  and the receiving section  25  shown in  FIG. 1 . In the present embodiment, a configuration of the transceiver is only different from the configuration of the transceiver in the first embodiment. 
     In the first embodiment, the device  2  receives the transmission signal two waves having the angular frequencies ω C1 +ω B1  and ω C1 −ω B1  from the device  1  and detects the phases θ H1 (t) and θ L1 (t) and the device  1  receives the transmission signal two waves having the angular frequencies ω C2 +ω B2  and ω C2 −ω B2  from the device  2  and detects the phases θ H2 (t) and θ L2 (t). The distance is calculated using these four phases. 
     However, in general, a band-pass filter (BPF) that suppresses frequency components other than a frequency component near a desired wave is inserted between the transceiver and the antenna. Since the filter is requested to have a steep frequency characteristic, in general, an order of the filter is high. Therefore, a delay of a signal passing through the filter is large. For example, a delay of approximately 10 ns occurs in a BPF in a 900 MHz band. If τ 1  increases by 10 ns because of a delay of the BPF of the transceiver, a calculated distance increases by 3 m. Moreover, a delay amount of the BPF has a temperature characteristic and involves an increase or a decrease of 10% or more according to an ambient environment. Therefore, unless a BPF delay time is not compensated during distance measurement, a distance cannot be accurately calculated. The present embodiment compensates for such a BPF delay time. 
     In  FIG. 16 , a multiplier TMIX 1  is equivalent to the multipliers TM 11  and TM 21  shown in  FIG. 4 or 5 . Similarly, multipliers TMIX 2  and RMIX 1  to RMIX 6  are equivalent to the multipliers TM 12  and  22  and the multipliers RM 11  and  21  to RM 16  and  26 . Adders TSUM 1 , RSUM 1 , and RSUM 2  are equivalent to the adders TS 11  and TS 21 , the adders RS 11  and RS 21 , and the adders RS 12  and RS 22 . 
     A transmitter including the multipliers TMIX 1  and TMIX 2  and the adder TSUM 1  and a receiver including the multipliers RMIX 1  to RMIX 6  and the adders RSUM 1  and RSUM 2  adopt the image suppression scheme like the transmitter and the receiver shown in  FIGS. 4 and 5 . In the present embodiment, multipliers RSMIX 1  and RSMIX 2  are provided in parallel to multipliers RMIX 1  and RMIX 2 . 
     An RF (high-frequency) input terminal RX 1  is connected to an RX terminal of a switch SW 1 . An RF output terminal TX 1  is connected to a TX terminal of the switch SW 1 . An FIL terminal of the switch SW 1  is connected to an input terminal of the band-pass filter BPF. A DET terminal of a switch SW 2  is connected to a common input terminal of an added RF input terminal sRX 1 , that is, a common input terminal of the multiplier RMIX 1  and the multiplier RMIX 2 . An FIL_I terminal of the switch SW 2  is coupled to an input terminal of the band-pass filter BPF by a not-shown coupler. An FIL_O terminal of the switch SW 2  is coupled to an output terminal of the band-pass filter BPF by a not-shown coupler. The output terminal of the band-pass filter BPF is connected to an antenna ANT terminal. Note that the switches SW 1  and SW 2  are controlled by the control section  11  (or the control section  21 ). 
     In the present embodiment, a delay time of the band-pass filter BPF (hereinafter referred to as BPF delay time) is measure in advance. The delay time is subtracted from a delay time calculated during phase detection to perform accurate distance measurement. The BPF delay time is obtained according to a difference between a signal delay time calculated by giving a transmission signal from the transmitter to the receiver via the switches SW 1  and SW 2  and a signal delay time calculated by giving the transmission signal to the receiver via the switch SW 1 , the band-pass filter BPF, and the switch SW 2 . 
     That is, when the BPF delay time is measured, the switch SW 1  selects the TX terminal and the switch SW 2  is connected to either one of the FIL_I terminal and the FIL_O terminal. In this state, the control sections  11  and  21  operate the transmitter. The receiver turns off only the multipliers RMIX 1  and RMIX 2  and operates the other circuit blocks. 
     The BPF delay time is calculated by detecting a transmission signal from the transmitter via the switch SW 2  and the multipliers RSMIX 1  and RSMIX 2 . The multiplier RMIX 1  and the multiplier RMIX 2  are turned off in order to prevent an error from increasing because a situation in which a signal leaks from the TX terminal to the RX terminal of the switch SW 1  and is added to a detection signal. If the multiplier RMIX 1  and the multiplier RMIX 2  are turned off, isolation of the multiplier RMIX 1  and the multiplier RMIX 2  is added to isolation of the switch SW 1 . Therefore, it is possible to greatly attenuate a signal directly leaking from the transmitter. 
     In order to detect the BPF delay time, the switch SW 2  is connected to the FIL_I terminal first to detect a signal via the FIL_I terminal and thereafter connected to the FIL_O terminal to detect a signal via the FIL_O terminal. Alternatively, the opposite sequence may be adopted. However, in the following explanation, the switch SW 2  is connected to the FIL_I terminal and thereafter connected to the FIL_O terminal. Note that the transmitter and the receiver that detect the BPF delay time are present in the same device and can use a common local signal. Therefore, there is no problem concerning an initial phase and a frequency shift. 
     When an angular frequency of a transmission wave is represented as ω C1 +ω B1  and an initial phase is represented as θ C1 +θ B1 , a signal of the TX 1  terminal is represented by the following Equation (169): 
         tx 1( t )=cos {(ω C1 +ω B1 ) t+θ   C1 +θ B1 }  (169)
 
     A signal delay due to the switch SW 1 , the not-shown couplers, and the switch SW 2  can be neglected. Therefore, the signal represented by Equation (169) is input to the multipliers RSMIX 1  and RSMIX 2 . The phase θ H1I (t) detected in this case is calculated by Equation (27) described above. That is, the phase θ H1I (t) is a result obtained by applying condition ω C1 =ω C2 , ω B1 =ω B2 , θ C1 =θ C2 , θ B1 =θ B2 , and θ τH1 =0. 
       θ H1I ( t )=0  (170)
 
     Similarly, when an angular frequency of a transmission wave is represented as ω C1 −ω B1  and an initial phase is represented as θ C1 −θ B1 , a signal of the TX 1  terminal is represented by the following Equation (171): 
         tx 1( t )=cos {(ω C1 −ω B1 ) t+θ   C1 −θ B1 }  (171)
 
     In this case, signals input to the multipliers RSMIX 1  and RSMIX 2  are the same as the signal represented by Equation (171). A phase θ L1I (t) detected in this case is calculated from Equation (47) described above. That is, the θ H1I (t) is a result obtained by applying conditions ω C1 =ω C2 , ω B1 =ω B2 , θ C1 =θ C2 , θ B1 =θ B2 , and θ τL1 =0. 
       θ L1I ( t )=0  (172)
 
     Subsequently, the switch SW 2  is connected to the FIL_O terminal. When a BPF delay is represented as τ BPF , signals having two frequencies input to the multipliers RSMIX 1  and RSMIX 2  via the switch SW 1 , the not-shown couplers, and the switch SW 2  are respectively represented by the following Equations (173) and (174): 
         rx 1( t )=cos {(ω C1 +ω B1 ) t+θ   C1 +θ B1 −θ BPFH }  (173)
 
         rx 1( t )=cos {(ω C1 −ω B1 ) t+θ   C1 −θ B1 − BPFL }  (174)
 
     In the equations, 
       θ BPFH =((ω C1 +ω B1 )τ BPF   (175)
 
       θ BPFL =((ω C1 +ω B1 )τ BPF   (176)
 
     From Equations (27) and (47) described above, phases θ H10 (t) and θ L10 (t) are represented by the following Equations (177) and (178): 
       θ H10 ( t )=θ BPFH   (177)
 
       θ L10 ( t )=θ BPFL   (178)
 
     The following Equation (179) is derived from Equations (170), (171), (177), and (178): 
       {θ H10 ( t )−θ L10 ( t )}−{θ H1I ( t )−θ L1I ( t )}=θ BPFH −θ BPFL =2ω B1 τ BPF   (179)
 
     A BPF delay time is calculated by Equation (179). The other components and action are the same as the components and the action in the first embodiment. That is, during actual use, the control sections  11  and  21  cause the switch SW 1  to select the TX terminal during transmission and select RX terminal during reception. 
     The control sections  11  and  21  correct τ 1  in the calculation of Equation (61) described above with the BPF delay time calculated from Equation (179) and thereafter calculate the distance R. 
     Note that, in the above explanation, the phase of the band-pass filter BPF input terminal is calculated assuming that the phase is equal to the TX output phase. However, if there is a delay in the switch SW, the coupler  1 , or the like, a phase difference is detected taking into account the delay. However, even during the actual use, a signal including a delay of the switch SW 1 , the coupler, or the like appear at a band-pass filter BPF output end and is supplied to the antenna ANT. Therefore, even if there is a delay before a band-pass filter BPF input end, it is possible to correctly perform measurement. Note that the control sections  11  and  21  may calculate a BPF delay time, for example, during a start of the actual use. 
     Incidentally, as an environment depending element, the band-pass filter BPF is mainly conceivable. Therefore, if a phase at the time when the switch SW 2  is connected to the FIL_I terminal, that is, a phase at the band-pass filter BPF input end less easily affected by an environment is saved in a memory in advance, there is no problem even if a value saved in the memory is used during subsequent calculation of a BPF delay time. That is, in this case, after the phase detection at the band-pass filter BPF input end, it is possible to always connect the switch SW 2  to the FIL_O terminal (the band-pass filter BPF output end). Consequently, it is possible to prevent a situation in which a power step change occurs and a large spurious occurs because the switch SW 2  is switched during transmission. 
     As explained above, in the present embodiment, effects same as the effects in the respective embodiments explained above. Further, there is an advantage that it is possible to perform accurate distance measurement by removing the influence of a delay time due to the band-pass filter. 
     Note that, in the present embodiment, in order to avoid a measurement error due to a lack of isolation of the switch SW 1 , the multiplier RSMIX 1  and the multiplier RSMIX 2  used during calculation of a BPF delay time are adopted. However, if sufficient isolation of the switch SW 1  can be secured, it is also possible to connect the DET terminal of the switch SW 2  to the RX 1  terminal and measure a delay via the multiplier RMIX 1  and the multiplier RMIX 2 . In this case, during reception, the switch SW 1  is connected to the RX 1  terminal and the switch SW 2  is connected to the FIL_I terminal or opened. In this case, the multiplier RSMIX 1  and the multiplier RSMIX 2  may be omitted. 
     Fourth Embodiment 
       FIGS. 17 and 18  are explanatory diagrams showing a fourth embodiment of the present invention. The present embodiment indicates an example in which the respective distance measuring systems are applied to a smart entry system. 
     In  FIG. 17 , a key  31  can transmit, by radio, a signal for enabling unlocking and locking of a door of an automobile  32  and a start of an engine of the automobile  32 . That is, the key  31  includes a not-shown data transmitting/receiving section and can transmit encrypted peculiar data for authentication with the data transmitting/receiving section. A radio wave from the data transmitting/receiving section of the key  31  is received in a not-shown vehicle control device  35  mounted on the automobile  32 . 
     As shown in  FIG. 18 , a control section  36  is provided in the vehicle control device  35 . The control section  36  controls respective sections of the vehicle control device  35 . The control section  36  is configured of a processor including a CPU. The control section  36  may operate a computer program stored in a memory  38  and control the respective sections. 
     A data transmitting/receiving section  37  is provided in the vehicle control device  35 . The data transmitting/receiving section  37  can perform wireless communication with the data transmitting/receiving section of the key  31  via an antenna  35   a . The data transmitting/receiving section  37  receives the peculiar data transmitted from the key  31  and transmits predetermined response data to the key  31  to perform authentication of the key  31  and the automobile  32 . 
     The data transmitting/receiving section  37  can finely set electric field intensity. The authentication is not performed unless the key  31  is located in a relatively close position where the key  31  is capable of receiving transmission data of the data transmitting/receiving section  37 , that is, near the automobile  32 . 
     For example, as indicated by a broken line in  FIG. 17 , it is assumed that the key  31  is located sufficiently close to the automobile  32 . In this case, the data transmitting/receiving section  37  is capable of communicating with the key  31 . The data transmitting/receiving section  37  authenticates the key  31  through collation with peculiar data recorded in a memory  37   a . The data transmitting/receiving section  37  outputs a signal indicating that the key  31  is authenticated to the control section  36 . Consequently, the control section  36  controls an unlocking/locking device  39  to give permission of locking or unlocking. 
     In  FIG. 17 , attackers of a relay attack carry relay devices  33  and  34 . The relay device  33  is capable of communicating with the key  31 . The relay device  34  is capable of communicating with the data transmitting/receiving section  37  in the automobile  32 . The relay devices  33  and  34  relay communication between the key  31  and the data transmitting/receiving section  37 . Consequently, even when the key  31  is sufficiently separated from the automobile  32  as shown in  FIG. 17  and direct communication between the key  31  and the data transmitting/receiving section  37  cannot be performed, the data transmitting/receiving section  37  can authenticate the key  31  through the relay devices  33  and  34 . 
     Therefore, in the present embodiment, the control section  36  determines on a basis of an authentication result of the data transmitting/receiving section  37  and a distance measurement result from the second device  2  whether unlocking and locking, a start of the engine, and the like are permitted. 
     The second device  2  in the respective embodiments is incorporated in the key  31 . On the other hand, the device  1  in the respective embodiments is mounted on the vehicle control device  35 . A transmission wave from the device  1  is received in the device  2  via an antenna  27   a . A transmission wave from the device  2  is received in the device  1  via the antenna  27   a . The transmission wave from the device  1  is directly received by the antenna  27   a  in some case and is received by the antenna  27   a  through the relay devices  33  and  34  in other cases. Similarly, the transmission wave from the second device  2  is directly received by the device  1  from the antenna  27   a  in some case and is received by the device  1  from the antenna  27   a  through the relay devices  33  and  34 . 
     When it is assumed that phases of the transmission waves from the device  1  and the device  2  do not change in the relay devices  33  and  34 , the device  2  can calculate a distance from the key  31  on a basis of the phases calculated in the devices  1  and  2 . The device  2  outputs the calculated distance to the control section  36 . A distance threshold for permitting authentication of the key  31  is stored in the memory  38 . When the distance calculated by the device  2  is within the distance threshold read out from the memory  38 , the control section  36  assumes that the key  31  is authenticated and permits unlocking and locking, a start of the engine, and the like. When the distance calculated by the device  2  is larger than the distance threshold read out from the memory  38 , the control section  36  does not permit the authentication of the key  31 . Therefore, in this case, the control section  36  does not permit unlocking and locking, a start of the engine, and the like. 
     Note that the relay devices  33  and  34  can change the phases of the transmission waves from the device  1  and the device  2 . Even in this case, since initial phases of the devices  1  and  2  are unknown, the relay devices  33  and  34  cannot calculate a phase shift amount necessary for keeping the distance calculated by the device  2  within the distance threshold read out from the memory  38 . Therefore, even if the relay devices  33  and  34  are used, possibility that the authentication of the key  31  is permitted is sufficiently small. 
     As explained above, in the present embodiment, by using the distance measuring system in the respective embodiments, it is possible to prevent unlocking and the like of a vehicle from being performed by a relay attack to the smart entry system. 
     (Transmission Sequence) 
       FIGS. 19 to 44  are explanatory diagrams showing various sequences adoptable in the respective embodiments. 
     (Examples of Two Waves) 
     First, various sequences for transmitting two waves from each of the devices  1  and  2  are explained with reference to  FIGS. 19 to 32 . 
     Two transmission waves output from the device  1  are represented as transmission waves A 1  and A 2 . Frequencies of the transmission waves A 1  and A 2  are respectively represented as f A1  and f A2 . Two transmission waves output from the device  2  are represented as transmission waves B 1  and B 2 . Frequencies of the transmission waves B 1  and B 2  are respectively represented as f B1  and f B2 . As explained above, taking into account the presence of a frequency band in which simultaneous transmission and reception is prohibited in the Radio Law, the transmission waves A 1 , A 2 , B 1 , and B 2  are transmitted in time-series processing in a predetermined sequence. 
       FIG. 19  shows a sequence in which carrier sense is performed, a response is absence, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS (second) and thereafter transmits the transmission wave A 1 . Further, the device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS and thereafter transmits the transmission wave A 2 . 
     The device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after a predetermined period after the transmission of the transmission wave A 2  from the device  1  without responding to reception of the transmission wave A 2  (a response is absent) and thereafter transmits the transmission wave B 1 . Further, the device  2  carries out the carrier sense at the frequency f B2  for, for example, 128 μS and thereafter transmits the transmission wave B 2 . Note that a phase detected in the device  2  may be reflected in phases of the transmission waves B 1  and B 2  in a predetermined period (which may include a carrier sense period) after the transmission of the transmission wave A 2  from the device  1 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in a carrier sense exception sequence explained below. 
       FIG. 20  shows a sequence in which the carrier sense is performed, a response is absent, and the repeated alternation is present. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . Further, the device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS without waiting for a response of a transmission wave from the device  2  and thereafter transmits the transmission wave A 2 . Further, the device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS without waiting for a response of a transmission wave from the device  2  and thereafter transmits the transmission wave A 1  again (the repeated alternation is present). 
     On the other hand, the device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after a predetermined period after the transmission of the transmission wave A 2  from the device  1  and thereafter transmits the transmission wave B 1 . Further, the device  2  carries out the carrier sense at the frequency f B2  for, for example, 128 μS without a response of a transmission wave from the device  1  and thereafter transmits the transmission wave B 2 . Further, the device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS without a response of a transmission wave from the device  1  and thereafter transmits the transmission wave B 1  again (the repeated alternation is present). 
     This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 21  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . 
     The device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS after a predetermined period after reception of the transmission wave B 1  from the device  2  and thereafter transmits the transmission wave A 2 . When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 22  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is present. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . 
     The device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS after a predetermined period after reception of the transmission wave B 1  from the device  2  and thereafter transmits the transmission wave A 2 . When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . 
     Further, in this example, the device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS after a predetermined period after reception of the transmission wave B 2  from the device  2  and thereafter transmits the transmission wave A 1  again (the repeated alternation is present). When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  again after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 23  shows a sequence in which the carrier sense is performed in some case and is not performed in other cases, a response is absent, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter continuously transmits the transmission waves A 1  and A 2 . 
     The device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after a predetermined period after reception of the transmission wave A 2  from the device  1  and thereafter continuously transmits the transmission waves B 1  and B 2 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense only at a start time of transmission. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 24  shows a sequence in which the carrier sense is performed in some case and is not performed in other cases, a response is present, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . When receiving the transmission wave B 1  from the device  2 , the device  1  transmits the transmission wave A 2  after, for example, 0 microsecond after the reception without carrying out the carrier sense. 
     When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense only at a start time of transmission. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 25  shows a sequence in which the carrier sense is absent, a response is absent, and the repeated alternation is absent. 
     The device  1  transmits the transmission wave A 1  without carrying out the carrier sense. Further, the device  1  transmits the transmission wave A 2  after several ten milliseconds after the transmission of the transmission wave A 1  without carrying out the carrier sense. The device  2  transmits the transmission wave B 1  after, for example, 100 milliseconds after reception of the transmission wave A 2  from the device  1  without carrying out the carrier sense. Further, the device  2  transmits the transmission wave B 2  after several ten milliseconds after the transmission of the transmission wave B 1  without carrying out the carrier sense. 
       FIG. 26  shows a sequence in which the carrier sense is absent, a response is absent, and the repeated alternation is absent. 
     The device  1  transmits the transmission wave A 1  without carrying out the carrier sense. Further, the device  1  once suspends the transmission after the transmission of the transmission wave A 1  and thereafter transmits the transmission wave A 2  without carrying out the carrier sense. In the figure, a transmission suspension section is described as 0 microsecond to 2 milliseconds. The device  2  immediately transmits the transmission wave B 1  after reception of the transmission wave A 2  from the device  1  without carrying out the carrier sense. In the figure, the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds after the reception of the transmission wave A 2  from the device  1 . Further, the device  2  once suspends transmission after the transmission of the transmission wave B 1  and thereafter transmits the transmission wave B 2  without carrying out the carrier sense. In the figure, a transmission suspension section is described as 0 microsecond to 2 milliseconds. 
       FIG. 27  shows a sequence in which the carrier sense is absent, a response is absent, and repeated alternation is present. 
     The device  1  transmits the transmission wave A 1  without carrying out the carrier sense. The device  1  transmits the transmission wave A 2  after several ten milliseconds after the transmission of the transmission wave A 1  without carrying out the carrier sense. Further, the device  1  transmits the transmission wave A 1  again after several ten milliseconds after the transmission of the transmission wave A 2  without carrying out the carrier sense. The device  2  transmits the transmission wave B 1  after, for example, 100 milliseconds after reception of the transmission wave A 1  from the device  1  without carrying out the carrier sense. Further, the device  2  transmits the transmission wave B 2  after, for example, 100 milliseconds after the transmission of the transmission wave B 1  without carrying out the carrier sense. Further, the device  2  transmits the transmission wave B 1  after, for example, 100 milliseconds after the transmission of the transmission wave B 2  without carrying out the carrier sense. 
       FIG. 28  shows a sequence in which the carrier sense is absent, a response is present, and the repeated alternation is absent. 
     The device  1  transmits the transmission wave A 1  without carrying out the carrier sense. When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . The device  1  transmits the transmission wave A 2  after several ten milliseconds after the transmission of the transmission wave A 1  without carrying out the carrier sense. When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . 
       FIG. 29  shows a sequence in which the carrier sense is absent, a response is present, and the repeated alternation is present. 
     The device  1  transmits the transmission wave A 1  without carrying out the carrier sense. When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . The device  1  transmits the transmission wave A 2  after several ten milliseconds after the transmission of the transmission wave A 1  without carrying out the carrier sense. When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . Further, the device  1  transmits the transmission wave A 1  again after several ten milliseconds after the transmission of the transmission wave A 2  without carrying out the carrier sense. When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  again after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . 
       FIG. 30  shows a sequence in which the carrier sense is performed in some case and is not performed in other cases, a response is present, and the repeated alternation is present. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . After reception of the transmission wave B 1  after several ten milliseconds after the transmission of the transmission wave A 1 , the device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS and thereafter transmits the transmission wave A 2 . When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . After the reception of the transmission wave B 1  after several ten milliseconds after the transmission of the transmission wave A 1 , the device  1  transmits the transmission wave A 1  again without carrying out the carrier sense this time. When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  again after, for example, 0 microsecond to 2 milliseconds after the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense only before the repeated alternation. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 31  shows a sequence in which the carrier sense is performed in some case and is not performed in other cases, a response is absent, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter continuously transmits the transmission waves A 1  and A 2 . The device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after reception of the transmission wave A 2  from the device  1  and thereafter continuously transmits the transmission waves B 1  and B 2 . Further, the device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after, for example, 100 milliseconds from the transmission of the transmission wave B 2  and thereafter continuously transmits the transmission waves B 1  and B 2  again. After receiving the transmission wave B 2  from the device  2 , the device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter continuously transmits the transmission waves A 1  and A 2 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense only at start times of respective transmissions. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 32  shows a sequence in which the carrier sense is performed in some case and is not performed in other cases, a response is present, and the repeated alternation is absent. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . After receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . After receiving the transmission wave B 1  from the device  2 , the device  1  transmits the transmission wave A 2  without carrying out the carrier sense. When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . 
     Further, the device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS after, for example, 100 milliseconds after the transmission of the transmission wave B 2  and transmits the transmission wave B 1 . When receiving the transmission wave B 1  from the device  2 , the device  1  transmits the transmission wave A 1  without carrying out the carrier sense in response to the reception of the transmission wave B 1 . After receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 2  without carrying out the carrier sense. When receiving the transmission wave B 2  from the device  2 , the device  1  transmits the transmission wave A 2  without carrying out the carrier sense in response to the reception of the transmission wave B 2 . Note that, in an idle time until the response, the device  2  may reflect a received and detected phase on phases of the transmission waves B 1  and B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense only at start times of the devices  1  and  2 . When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 33  shows a sequence in which the carrier sense is present, a response is present, and the repeated alternation is present. 
     The device  1  carries out the carrier sense at the frequency f A1  for, for example, 128 μS and thereafter transmits the transmission wave A 1 . When receiving the transmission wave A 1  from the device  1 , the device  2  transmits the transmission wave B 1  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 1 . After the transmission of the transmission wave B 1 , the device  2  carries out the carrier sense at the frequency f B1  for, for example, 128 μS and thereafter transmits the transmission wave B 1 . When receiving the transmission wave B 1  from the device  2 , the device  1  transmits the transmission wave A 1  without carrying out the carrier sense in response to the reception of the transmission wave B 1 . Further, after transmitting the transmission wave A 1 , the device  1  carries out the carrier sense at the frequency f A2  for, for example, 128 μS and thereafter transmits the transmission wave A 2 . When receiving the transmission wave A 2  from the device  1 , the device  2  transmits the transmission wave B 2  after, for example, 0 microsecond to 2 milliseconds from the reception without carrying out the carrier sense in response to the reception of the transmission wave A 2 . Further, after the transmission of the transmission wave B 2 , the device  2  carries out the carrier sense at the frequency f B2  for, for example, 128 μS and thereafter transmits the transmission wave B 2 . When receiving the transmission wave B 2  from the device  2 , the device  1  transmits the transmission wave A 2  without carrying out the carrier sense in response to the reception of the transmission wave B 2 . 
     This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
     (Examples of Three Waves) 
     Various sequences for transmitting three waves from each of the devices  1  and  2  are explained with reference to  FIGS. 34 to 44 . 
     Three transmission waves output from the device  1  are represented as transmission waves A 1 , A 2 , and A 3 . Frequencies of the transmission waves A 1 , A 2 , and A 3  are respectively represented as f A1 , f A2 , and f A3  Three transmission waves output from the device  2  are represented as transmission waves B 1 , B 2 , and B 3 . Frequencies of the transmission waves B 1 , B 2 , and B 3  are respectively represented as f B1 , f B2 , and f B3 . As explained above, taking into account the presence of a frequency band in which simultaneous transmission and reception is prohibited in the Radio Law, the transmission waves A 1  to A 3  and B 1  to B 3  are transmitted in time-series processing in a predetermined sequence. 
     Note that a method of describing sequences of transmission and reception in  FIGS. 34 to 44  is the same as the method shown in  FIGS. 19 to 32 . In the following explanation, explanation is omitted concerning specific sequences of transmission and reception. 
       FIG. 34  shows a sequence in which the carrier sense is performed, a response is absent, and the repeated alternation is absent. This example of the sequence is an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 35  shows a sequence in which the carrier sense is performed, a response is absent, and the repeated alternation is present. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 36  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is absent. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 37  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is present. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 38  shows a sequence in which the carrier sense is absent, a response is absent, and the repeated alternation is absent. 
       FIG. 39  shows a sequence in which the carrier sense is absent, a response is present, and the repeated alternation is present. 
       FIG. 40  shows a sequence in which the carrier sense is absent, a response is present, and the repeated alternation is absent. 
       FIG. 41  shows a sequence in which the carrier sense is absent, a response is present, and the repeated alternation is present. 
       FIG. 42  shows a sequence in which the carrier sense is present in some case and is absent in other cases, a response is present, and the repeated alternation is absent. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 43  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is present. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
       FIG. 44  shows a sequence in which the carrier sense is performed, a response is present, and the repeated alternation is present. This example of the sequence is also an example in which presence or absence of a distance measurement frequency is determined by the carrier sense. When the distance measurement frequency is sensed by the carrier sense, (a) the sequence may be started from the beginning again after a predetermined time (e.g., several milliseconds) or (b) may be started in the carrier sense exception sequence. 
     (Modification) 
       FIGS. 45 to 47  show a modification. In the modification, a period in which an initial phase should be fixed is explained.  FIG. 45  is an explanatory diagram showing a relation between a transmission sequence and a period in which an initial phase is maintained.  FIG. 46  is an explanatory diagram showing a carrier frequency used for distance measurement.  FIG. 47  is a flowchart for explaining the modification. 
     In the first embodiment, the simultaneous transmission and the simultaneous reception of the respective two frequencies from the devices  1  and  2  are assumed. In a period of the transmission and reception, the oscillators  13  and  23  are caused to continue oscillation such that the initial phase does not change. On the other hand, in the second embodiment, between the devices  1  and  2 , it is specified that only one wave can be transmitted and received at the same time, transmission and reception of at least four waves necessary for distance measurement is carried out in time-series processing, and carrier signals from the devices  1  and  2  are repeatedly alternated to enable accurate distance measurement even when the time-series transmission and reception is performed. 
     For example, as explained above, in the example shown in  FIG. 15 , the transmission and reception of the transmission wave having the angular frequency ω C1 +ω B1  from the device  1 , the two times of transmission and reception of the transmission wave having the angular frequency ω C2 +ω B2  from the device  2 , and the transmission and reception of the transmission wave having the angular frequency ω C1 +ω B1  from the device  1  are performed. 
     Further, the transmission and reception of the transmission wave having the angular frequency ω C1 −ω B1  from the device  1 , the two times of transmission and reception of the transmission wave having the angular frequency ω C2 −ω B2  from the device  2 , and the transmission and reception of the transmission wave having the angular frequency ω C1 −ω B1  from the device  1  are performed. 
     An addition result of phases obtained in the devices  1  and  2  according to first transmission and reception of four waves in  FIG. 15  is shown in a former half portion (first to fourth terms) of Equation (139) described above. The addition result of the first to fourth terms is 2(θ τH1 +θ τH2 ) as shown in Equation (132) described above. As shown in Equations (20) and (30) described above, the addition result does not include a term of an initial phase. That is, information concerning the initial phase is not included in an operation result of phases obtained by the first transmission and reception of the four waves in  FIG. 15 . Therefore, in an operation of the first to fourth terms of Equation (139) described above, a correct result is obtained if the initial phase is not changed only in the transmission and reception period of the four waves. 
     Similarly, an addition result of phases obtained in the devices  1  and  2  by last transmission and reception of fourth waves in  FIG. 15  is shown in a latter half portion (fifth to eighth terms) of Equation (139) described above. The addition result of the fifth to eighth terms is 2(θ τL1 +θ τL2 ) as shown in Equation (138) described above. As shown in Equations (40) and (50) described above, the addition result does not include a term of an initial phase. That is, information concerning the initial phase is not included in an operation result of phases obtained by last transmission and reception of the four waves in  FIG. 15 . Therefore, in an operation of the fifth to eighth terms of Equation (139) described above, a correct result is obtained if the initial phase is not changed only in the transmission and reception period of the four waves. 
       FIG. 45  shows this state and indicates that an initial phase is maintained fixed in a distance measurement period in which the angular frequencies ω C1 −ω B1  and ω C2 +ω B2  are used and an initial phase is maintained fixed in a distance measurement period in which the angular frequencies ω C1 −ω B1  and θ C2 −ω B2  are used. 
     That is, for example, when the sequence of  FIG. 15  is adopted, it is sufficient to cause the oscillators  13  and  23  to continue oscillation such that the initial phase does not change in a period of the first transmission and reception of the four waves and cause the oscillators  13  and  23  to continue oscillation such that the initial phase does not change in a period of last transmission and reception of the four waves. Even if the oscillation of the oscillators  13  and  23  stops and the initial phase changes until transmission and reception of a fifth wave from transmission and reception of a fourth wave, an accurate distance can be calculated on a basis of Equation (139) described above. 
     On the other hand, even if the oscillators  13  and  23  continuously operate, the initial phase changes if a frequency setting is not changed in the ω C +ω B  distance measurement section and if a frequency setting is not changed in the ω C −ω B  distance measurement section. This is explained in detail below (concerning two carrier signals). 
     Incidentally, in the respective embodiments, the example is explained in which the two carrier signals transmitted by the devices  1  and  2  has a frequency of a sum of or a difference between, for example, the relatively high angular frequencies ω C1  and ω C2  and, for example, the relatively low angular frequencies ω B1  and ω B2  Note that the angular frequencies ω C1  and ω C2  are set to substantially the same frequency and the angular frequencies ω B1  and ω B2  are set to substantially the same frequency. 
     (Concerning Two Carrier Signals) 
     However, in the distance measurement in the respective embodiments, as explained below, the devices  1  and  2  only have to transmit two carrier signals respectively having predetermined frequency differences. 
     An angular frequency ω C +ω B  and an angular frequency ω C −ω B  can be modified as described below. 
       ω C +ω B =(ω C −Δω c )+(Δω C +ω B )=ω′ C +ω H   (201)
 
       ω C −ω B =(ω C −Δω c )+(Δω C −ω B )=ω′ C +ω L   (202)
 
     When transmission waves having the angular frequency ω C  and the angular frequency ω B  are represented as f C  and f B  on a frequency axis, transmission waves having the angular frequency ω C +ω B  and the angular frequency ω C −ω B  are as shown on a left side of  FIG. 46 . 
     Equations (201) and (202) described above indicate that a representation method is changed without changing a frequency. The transmission wave having the angular frequency ω C +ω B  is obtained by a sum of transmission waves having an angular frequency ω′ C  and an angular frequency ω H . The transmission wave having the angular frequency ω C −ω B  is obtained by a sum of transmission waves having the angular frequency ω′ C  and an angular frequency ω L . 
     The center in  FIG. 46  shows transmission waves based on this representation. The transmission waves having the angular frequencies ω′ C , ω H , and ω L , are respectively represented as f C , f H , and f L  on the frequency axis. A transmission wave having a frequency f C −f B  and a transmission wave having a frequency f C +f B  on the left side of  FIG. 46  respective indicates that the transmission waves can be represented as a transmission wave having the same frequency f C +f L  and a transmission wave having the same frequency f C +f H . 
     That is, Equations (201) and (202) described above indicate that two carriers transmitted by the devices  1  and  2  do not need to be obtained by a sum of and a difference between two frequencies and the devices  1  and  2  only have to generate two carriers having a predetermined frequency difference and transmit the carriers. 
     As an example in which the transmission waves having the angular frequency ω C +ω B  and the angular frequency ω C −ω B  shown on the left side of  FIG. 46  are obtained, for example, the oscillators  13  and  23  shown in  FIG. 1  only have to generate a local signal having the angular frequency we as a relatively high local frequency (hereinafter referred to as RF-LO frequency) and a local signal having the angular frequency ω B  as a relatively low local frequency (hereinafter referred to as IF-LO frequency) and the transmitting section  14  and  24  only have to generate signals having a sum of or a difference between the local frequencies as transmission signals. 
     On the other hand, the transmission waves in the center of  FIG. 46  are obtained by generating a local signal having the angular frequency ω′ C  as the RF-LO frequency and generating two local signals having the angular frequencies ω L  and ω H  as the IF-LO frequency and generating, in the transmitting sections  14  and  24 , signals having a sum of these local frequencies as transmission signals. That is, the transmission waves in the center of  FIG. 46  are obtained by fixing the RF-LO frequency and changing the IF-LO frequency. 
     The angular frequency ω C −ω B  can be modified as shown below. 
       ω C −ω B =(ω C −2ω B )+(2ω B −ω B )=ω″ C +ω B   (203)
 
     A right side of  FIG. 46  shows transmission waves based on the representation of Equation (203) and indicates that, when a transmission wave having an angular frequency ω″ C  is represented as f″ C  on the frequency axis, the transmission wave having the frequency f C −f B  on the left side of  FIG. 46  can be represented as a transmission wave having the same frequency f″ C +f B . 
     For example, the transmission waves on the right side of  FIG. 46  are obtained by generating a local signal having the angular frequency ω′ C  and a local signal having the angular frequencies ω″ C  as the RF-LO frequency and generating a local signal having the angular frequency ω B  as the IF-LO frequency and generating, in the transmitting sections  14  and  24 , signals having a sum of these local frequencies as transmission signals. That is, the transmission waves on the right side of  FIG. 46  are obtained by fixing the IF-LO frequency and changing the RF-LO frequency. 
     (Change of the Carrier Frequencies) 
     In this way, the devices  1  and  2  only have to transmit the two carrier signals respectively having the predetermined frequency difference in the distance measurement. Moreover, when transmission and reception of a carrier having a higher frequency of the two carriers is alternately performed and transmission and reception of a carrier having a low frequency is subsequently alternately performed as in the sequence shown in  FIG. 15 , initial phases may be different in the transmission and reception of the carrier having the high frequency and the transmission and reception of the carrier having the low frequency. 
     Further, only the carrier having the high frequency is used for an operation of Equation (132) described above obtained by decomposing Equation (139) described above and only the carrier having the low frequency is used for an operation of Equation (138) described above. That is, the operation of Equation (132) described above and the operation of Equation (138) described above only have to be independently performed. The local frequencies may be changed between a transmission and reception period of the carrier having the high frequency and a transmission and reception period of the carrier having the low frequency. 
     When the change in the initial phase and the change in the carrier angular frequency are allowed, a flow shown in  FIG. 47  can be adopted instead of the flow shown in  FIG. 11A . 
     In the flow shown in  FIG. 47 , the device  1  sets a first local (LO) frequency before one wave transmission signal generation and the device  2  sets a first local (LO) frequency before reception of one wave transmission wave from the device  1 . The device  1  generates a carrier having a high frequency, for example, as one wave transmission signal using the local signal having the first LO frequency. The device  2  receives the one wave transmission wave using the carrier having the high frequency and acquires I and Q signals. 
     After acquisition of the I and Q signals based on the transmission signal having the high frequency from the device  2  and before generation of the next one wave transmission signal, the device  1  sets a second local (LO) frequency. Similarly, after the transmission of the one wave transmission wave to the device  1  and before reception of the next one wave transmission wave, the device  2  sets a second local (LO) frequency. The device  1  generates a carrier having a low frequency, for example, as one wave transmission signal using the local signal having the second LO frequency. The device  2  receives the transmission wave of the device  1  using the carrier having the low frequency and acquires I and Q signals. 
     (Configuration Example of the Oscillators and the Transceivers) 
     In this way, the devices  1  and  2  only have to be capable of generating and transmitting the two carrier signals having the predetermined frequency difference. Circuits having various configurations can be adopted as the oscillators  13  and  23 , the transmitting sections  14  and  24 , and the receiving sections  15  and  25  in  FIG. 1 . 
       FIG. 48A  is an explanatory diagram showing, in a simplified manner, an example of the configurations of the oscillator  13 , the transmitting section  14 , and the receiving section  15  of the device  1 .  FIG. 48B  is an explanatory diagram showing, in a simplified manner, an example of the configurations of the oscillator  23 , the transmitting section  24 , and the receiving section  25  of the device  2 . 
     As shown in  FIGS. 48A and 48B , the device  1  and the device  2  include an oscillator that generates an IF-LO frequency and an oscillator that generates an RF-LO frequency. Oscillation frequencies of the oscillators are, for example, fixed. Two carrier signals having a frequency difference can be generated by adding the IF-LO frequency to or subtracting the IF-LO frequency from the RF-LO frequency. 
       FIG. 49A  is an explanatory diagram showing, in a simplified manner, an example of the configurations of the oscillator  13 , the transmitting section  14 , and the receiving section  15  of the device  1 .  FIG. 49B  is an explanatory diagram showing, in a simplifier manner, an example of the configurations of the oscillator  23 , the transmitting section  24 , and the receiving section  25  of the device  2 . 
     The examples shown in  FIGS. 49A and 49B  are examples in which two carrier signals having a frequency difference are generated by an oscillator having a variable frequency and a frequency divider (N-div). 
       FIG. 50  is a circuit diagram more specifically showing an example of a circuit that generates signals given to the multipliers TM 11  and TM 12  in  FIG. 4 .  FIG. 51  is a circuit diagram more specifically showing an example of a circuit that generates signals given to the multipliers TM 21  and TM 22  in  FIG. 5 . 
     In  FIG. 50 , a multiplier TM 15  gives, to an adder TS 15 , a multiplication result of an I T  signal and a local signal cos(ω B1 t+ω B1 ) from the oscillator  13 . A multiplier TM 16  gives, to the adder TS 15 , a multiplication result of a Q T  signal and a local signal ±sin(ω B1 t+θ B1 ) from the oscillator  13 . The adder TS 15  adds up the two inputs and gives an addition result to the multiplier TM 11 . 
     A multiplier TM 17  gives, to an adder TS 16 , a multiplication result of the I T  signal and the local signal ±sin(ω B1 t+θ B1 ) from the oscillator  13 . A multiplier TM 18  gives, to the adder TS 16 , a multiplication result of the Q T  signal and the local signal cos(ω B1 t+θ B1 ) from the oscillator  13 . The adder TS 16  subtracts the output of a multiplier TM 18  from the output of the multiplier TM 17  and gives a subtraction result to the multiplier TM 12 . The other components are the same as the components shown in  FIG. 4 . 
     In  FIG. 51 , a multiplier TM 25  gives, to an adder TS 25 , a multiplication result of the I T  signal and a local signal cos(ω B2 t+θ B2 ) from the oscillator  23 . A multiplier TM 26  gives, to the adder TS 25 , a multiplication result of the Q T  signal and a local signal ±sin(ω B2 t+θ B2 ) from the oscillator  23 . The adder TS 25  adds up the two inputs and gives an addition result to the multiplier TM 21 . 
     A multiplier TM 27  gives, to an adder TS 26 , a multiplication result of the I T  signal and the local signal ±sin(ω B2 t+θ B2 ) from the oscillator  23 . A multiplier TM 28  gives, to the adder TS 26 , a multiplication result of the Q T  signal and the local signal cos(ω B2 t+θ B2 ) from the oscillator  23 . The adder TS 26  subtracts the output of the multiplier TM 28  from the output of the multiplier TM 27  and gives a subtraction result to the multiplier TM 22 . The other components are the same as the components shown in  FIG. 5 . 
       FIG. 52  is a circuit diagram showing an example of specific configurations of the transmitting section  14  and the receiving section  15  shown in  FIG. 1 .  FIG. 53  is a circuit diagram showing an example of specific configurations of the transmitting section  24  and the receiving section  25  shown in  FIG. 1 . Note that  FIGS. 52 and 53  show transceivers having a heterodyne configuration. 
     In  FIG. 52 , a multiplier TM 1 A gives, to an adder TS 1 A, a multiplication result of the I T  signal and the local signal cos(ω B1 t+θ B1 ) from the oscillator  13 . A multiplier TM 1 B gives, to the adder TS 1 A, a multiplication result of the Q T  signal and the local signal sin(ω B1 t+θ B1 ) from the oscillator  13 . The adder TS 1 A subtracts the output of the multiplier TM 1 B from the output of the multiplier TM 1 A and gives a subtraction result to a multiplier TM 1 C. The multiplier TM 1 C multiplies together the output of the adder TS 1 A and a local signal cos(ω C1 t+θ C1 ) and outputs a multiplication result as the transmission signal tx 1 . 
     A multiplier RM 1 A multiplies together the received signal rx 1  and the local signal cos(ω C1 t+θ C1 ) to obtain an I 1  signal and outputs the I 1  signal to multipliers RM 1 B and RM 1 C. The multiplier RM 1 B outputs a multiplication result of the I 1  signal and the local signal cos(ω B1 t+θ B1 ) as an I signal. The multiplier RM 1 C outputs a multiplication result of the I 1  signal and the local signal sin(ω B1 t+θ B1 ) as a Q signal. 
     In  FIG. 53 , a multiplier TM 2 A gives, to an adder TS 2 A, a multiplication result of the I T  signal and the local signal cos(ω B2 t+θB 2 ) from the oscillator  23 . A multiplier TM 2 B gives, to the adder TS 2 A, a multiplication result of the Q T  signal and the local signal sin(ω B2 t+θ B2 ) from the oscillator  23 . The adder TS 2 A subtracts the output of the multiplier TM 2 B from the output of the multiplier TM 2 A and gives a subtraction result to a multiplier TM 2 C. The multiplier TM 2 C multiplies together the output of the adder TS 2 A and a local signal cos(ω C2 t+ω C2 ) and outputs a multiplication result as the transmission signal tx 2 . 
     A multiplier RM 2 A multiplies together the received signal rx 2  and the local signal cos(ω c2 t+ω C2 ) to obtain the I 1  signal and outputs the I 1  signal to multipliers RM 2 B and RM 2 C. The multiplier RM 2 B outputs a multiplication result of the I 1  signal and the local signal cos(ω B2 t-ω B2 ) as the I signal. The multiplier RM 2 C outputs a multiplication result of the I 1  signal and the local signal sin(ω B2 t+θ B2 ) as the Q signal. 
       FIG. 54  is a circuit diagram showing an example of the specific configurations of the transmitting section  14  and the receiving section  15  shown in  FIG. 1 .  FIG. 55  is a circuit diagram showing an example of the specific configurations of the transmitting section  24  and the receiving section  25  shown in  FIG. 1 . Note that  FIGS. 54 and 55  show transceivers in which a direct conversion scheme is adopted. 
     In  FIG. 54 , a multiplier TM 1 D gives, to an adder TS 1 C, a multiplication result of the I T  signal and the local signal cos(ω C1 t+θ C1 ) from the oscillator  13 . A multiplier TM 1 E gives, to the adder TS 1 C, a multiplication result of the Q T  signal and the local signal sin(ω C1 t+θ C1 ) from the oscillator  13 . The adder TS 1 C subtracts the output of the multiplier TM 1 E from the output of the multiplier TM 1 D and outputs a subtraction result as the transmission signal tx 1 . 
     A multiplier RM 1 D multiplies together the received signal rx 1  and the local signal cos(ω C1 t+θ C1 ) and outputs a multiplication result as the I signal. A multiplier RM 1 E multiplies together the received signal rx 1  and the local signal sin(ω C1 t+θ C1 ) and outputs a multiplication result as the Q signal. 
     In  FIG. 55 , a multiplier TM 2 D gives, to the adder TS 2 C, a multiplication result of the I T  signal and the local signal cos(ω C2 t+θ C2 ) from the oscillator  23 . A multiplier TM 2 E gives, to the adder TS 2 C, a multiplication result of the Q T  signal and the local signal sin(ω C2 t+θ C2 ) from the oscillator  23 . The adder TS 2 C subtracts the output of the multiplier TM 2 E from the output of the multiplier TM 2 D and outputs a subtraction result as the transmission signal tx 2 . 
     A multiplier RM 2 D multiplies together the received signal rx 2  and the local signal cos(ω C2 t+θ C2 ) and outputs a multiplication result as the I signal. A multiplier RM 2 E multiplies together the received signal rx 2  and the local signal sin(ω C2 t+θ C2 ) and outputs a multiplication result as the Q signal. 
     (Transmission Example of Phase Information) 
     In the respective embodiments, the phase information is transmitted from either one of the first device and the second device to the other. However, as explained above, a method of transmitting the phase information is not particularly limited. For example, the phase information may be transmitted by shifting by a phase obtained from a received signal, a phase of a carrier signal to be transmitted. 
     For example, in this case, it is possible to adopt a flow in which the broken line portion of  FIG. 6  is replaced with a flow shown in  FIG. 56  and steps S 7 , S 8 , and S 18  in  FIG. 6  are omitted. 
       FIG. 56  is a flowchart for explaining an example corresponding to  FIG. 11A  in which the second device transmits phase information to the first device.  FIG. 56  includes a step of adding up a received phase and an initial phase between the acquisition step for the I and Q signals and the one wave transmission signal generating step in the second device shown in  FIG. 11A . Consequently, a carrier signal in which a phase detected from a received signal is added to the initial phase of the device  2  is generated. The device  1  can acquire phase information calculated by the device  2  by receiving the carrier signal from the device  2  and calculating a phase. 
     As extension of the addition of the received phase and the initial phase, for example, in the eight-times repeated alternating sequence shown in  FIG. 15 , a sequence of θ L2 , θ L2 , and θ L1  is considered, that is, after the device  2  transmits signals to the device  1  twice, the device  1  transmits a signal to the device  2 . A received phase added to the initial phase during last θ L1  transmission may be a phase of the immediately preceding θ L2  but may be a value obtained by taking into account a phase detection result of θ L2  immediately preceding θ 1,2 . That is, in addition to simply reflecting an immediately preceding received result, a result reflecting phases in the past may be added. In this case, in general, a result obtained by performing a predetermined operation on detected phase in the past is added to the initial phase. Further, in the case of the sequence shown in  FIG. 15 , since the device  2  receives a signal last, the device  2  is a device that calculates a distance. In this case, the device  1  does not always need to calculate a distance. 
       FIG. 57  is a flowchart for explaining an example corresponding to  FIG. 47 . A flow shown in  FIG. 57  includes a step of adding up a received phase and an initial phase between the acquisition step for I and Q signals and the one wave transmission signal generating step in the device  2  shown in  FIG. 47 . When the eight-time repeated alternating sequence shown in  FIG. 15  is assumed, as explained above, a predetermined calculation result with respect to detected phases in the past is added to the initial phase. Further, the device  2  is a device that calculates a distance. In this way, even when the carrier frequency is changed halfway in the distance measurement, the phase information may be transmitted while being included in the information concerning the phase of the carrier signal transmitted for the distance measurement. 
     (Doubling of a Measurement Distance) 
     Incidentally, the result of Equation (139) described above is a value in a system of residue of 2π, that is, a value between 0 and 2π. The distance R is R=τ 1 /C. When a left side of Equation (139) described above is represented as S A  and R is calculated, the following Equation (210) is obtained: 
         R =( S   A /4)×{ c /(ω B1 +ω B2 )}  (210)
 
     A maximum of S A  of Equation (210) described above is 2π. When (ω B1 +ω B2 )=2πΔf, a maximum R MAX  of the distance R is represented by the following Equation (211): 
         R   MAX   =c/ 4Δ f   (211)
 
     That is, Equation (211) described above indicates a maximum distance measurable without adopting the methods shown in  FIGS. 9 and 10  in the sequence shown in  FIG. 15  for transmitting two carriers having different frequencies four times from each of the device  1  and the device  2 , eight times in total (hereinafter referred to as eight-time alternate sequence). On the other hand, when four waves are simultaneously transmitted and a distance is calculated by Equation (61) described above, R MAX =c/2Δf. That is, in the eight-time alternate sequence shown in  FIG. 15 , a measurable distance is a half of a measurable distance during the four-wave simultaneous transmission. 
     Therefore, a method of increasing the measurable distance in the eight-time alternate sequence shown in  FIG. 15  is examined. When ω B1 −ω B2  represents a beat angular frequency and a result obtained by multiplying the time t 0  with a double of the beat angular frequency is represented as A(=2(ω B1 −ω B2 )t 0 ). A can be calculated by the following Equation (212) in which measured eight phases are used: 
         A =[mod {(θ H1 ( t )+θ H2 ( t+t   0 )−θ H1 ( t+t   0   +D )−θ H2 ( t+D )−(θ L1 ( t+T )+θ L2 ( t+t   0   +T )−θ L1 ( t+t   0   +D+T )−θ L2 ( t+D+T ))+π,2π}−π]/2  (212)
 
     When the following Equation (213) is satisfied, Equation (129) is modified to represent a distance with the following Equation (214): 
       |2(ω B1 −ω B2 ) t   0 |≤π/2  (213)
 
         R =[{θ H1 ( t )+θ H2 ( t+t   0 )−(θ L1 ( t+T )+θ L2 ( t+t   0   +T ))− A}/ 2]× c /(ω B1 +ω B2 )  (214)
 
     Therefore, the maximum R MAX  of the distance R in this case is represented by the following Equation (215): 
         R   MAX   =c/ 2Δ f   (215)
 
     That is, by performing distance measurement using Equation (214) described above, in the eight-time alternate sequence shown in  FIG. 15 , it is possible to set a transmittable and measurable maximum distance to a measurement distance same as the measurement distance during the four-wave simultaneous transmission without adopting the methods shown in  FIGS. 9 and 10 . 
     For example, if ω B1 =2π×5 [MHz], an error of ω B2  with respect to ω B1  is 40 [ppm], and t 0 =100 [μs], a left side of Equation (213) described above is 2(ω B1 −ω B2 )t 0 =2×2π×5×40×10 −6 ×100=0.08π. It is relatively easy to perform system design to satisfy Equation (213) described above. 
     [Notes] 
     [Note 1] 
     A distance measuring device that calculates a distance on a basis of carrier phase detection, the distance measuring device including a calculating section configured to calculate, on a basis of phase information acquired by a first device and a second device, at least one of which is movable, a distance between the first device and the second device, wherein 
     the first device includes: 
     a first reference signal source; and 
     a first transceiver configured to transmit two or more first carrier signals and receives two or more second carrier signals using an output of the first reference signal source, 
     the second device includes: 
     a second reference signal source configured to operate independently from the first reference signal source; and 
     a second transceiver configured to transmit the two or more second carrier signals and receives the two or more first carrier signals using an output of the second reference signal source, and 
     the calculating section calculates the distance on a basis of a phase detection result obtained by reception of the first and second carrier signals. 
     [Note 2] 
     The distance measuring device according to the note 1, wherein the calculating section calculates the distance by calculating a residue of 2π concerning an addition result of a residue result of 2π concerning the first phase difference and a residue result of 2π concerning the second phase difference. 
     [Note 3] 
     The distance measuring device according to the note 1, wherein 
     the first and second reference signal sources generate two kinds of local signals, and 
     the first and second transceivers are configured of a wireless receiver of an image suppression scheme in which the two kinds of local signals are used. 
     [Note 4] 
     The distance measuring device according to the note 1, further including: 
     a band-pass filter provided between the first transceiver and an antenna; and 
     a switch circuit that switches a first route for giving an output of a transmitter in the first transceiver to a receiver in the first transceiver via the band-pass filter and a second route for giving the output to the receiver in the first transceiver not via the band-pass filter, wherein 
     a receiver of the first transceiver includes two inputs, one of which is connected to an output of the switch circuit, and 
     the calculating section calculates a delay time due to the band-pass filter on a basis of a phase of the first carrier signal that passes the first route and a phase of the first carrier signal that passes the second route. 
     [Note 5] 
     The distance measuring device according to the note 1, wherein 
     the first carrier signal is two or more carrier signals having different frequencies, 
     the second carrier signal is two or more carrier signals having frequencies respectively corresponding to the two or more carrier signals of the first carrier signal, and 
     the first and second transceivers do not change initial phases and frequencies of the first and second carrier signals during a period in which carrier signals having frequencies corresponding to each other of the first and second carrier signals are transmitted and received. 
     [Note 6] 
     The distance measuring device according to the note 1, wherein one device of the first and second devices generates a carrier signal obtained by adding, to an initial phase, a phase detection result obtained by reception of a carrier signal from another device of the first and second devices and transmits the carrier signal to the other device, and the device that receives the carrier signal last in a distance measurement sequence is a device that performs distance measurement calculation. 
     While certain embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the inventions. Indeed, the novel devices and methods described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the embodiments described herein may be made without departing from the spirit of the inventions. The accompanying claims and their equivalents are intended to cover such forms or modification as would fall within the scope and spirit of the inventions.