Patent Publication Number: US-6982890-B2

Title: Three phase isolated vector switching AC to AC frequency converters

Description:
REFERENCE TO GOVERNMENT RIGHTS 
   This invention was made with United States government support awarded by the following agency: NAVY/ONR N0014-01-1-0623. The United States government has certain rights in this invention. 

   FIELD OF THE INVENTION 
   The present invention relates generally to the field of electrical power conversion and particularly to variable input and/or output frequency AC to AC power converters. 
   BACKGROUND OF THE INVENTION 
   Many power conversion applications require the conversion of AC power at one frequency to AC power at a higher or lower frequency. One common utilization for such power converters is the variable speed control of AC motors. The most common commercial AC to AC static switch frequency converters utilize an intermediate DC stage. One type of commercial converter, illustrated schematically in  FIG. 1 , utilizes a rectification stage  20  that provides DC current through an inductor  21  to an inverter stage  22  composed a multiple static switches  23 , which are illustrated in  FIG. 2 . A second type of AC-AC-converter, illustrated schematically in  FIG. 3 , has a rectification stage  28  that provides DC voltage on DC bus lines  29  and  30  to an inverter stage  32  composed of static switches  33 , as illustrated, for example, in  FIG. 4 . An energy storage capacitor  34  is connected across the DC bus lines  29  and  30 . While the DC link configuration of  FIG. 3  is widely used in commercial power converters, the DC link capacitor  34  constitutes one life-limiting component in these types of converters, as well as contributing to the bulk and cost of the converter. As an alternative to power conversion systems having an intermediate DC link, a variety of direct AC to AC converters have been developed. An example of a prior AC to AC converter is the matrix converter, shown schematically in  FIG. 5 , which utilizes one pole, three throw switches  36  to directly convert an AC input voltage at one frequency to an AC output voltage at another frequency. Matrix converters require bidirectional high power semiconductor switches, which are not presently commercially available as single units, but which can be implemented utilizing back to back IGBTs (insulated gate bipolar transistors)  37  and diodes as shown in  FIG. 6 . Because of the relatively high currents and voltages which these switches must handle, the semiconductor switches required are relatively expensive and can limit the reliability of the converter system. 
   SUMMARY OF THE INVENTION 
   In accordance with the present invention, an AC to AC frequency converter system includes a three-phase isolation transformer having three-phase input terminals and multiple sets of three-phase output terminals. The transformer provides multiple sets of three-phase output voltages at secondary output terminals. The transformer may be constructed to provide three or more sets of output voltages at the secondary output terminals, with four sets being preferred. Where four sets of output terminals are utilized, the three-phase voltages at each set of secondary output terminals are one or more multiples of 90° out of phase with the voltages on the other sets of output terminals. More generally, where the number of sets of output terminals is n, the three-phase voltages at each set of output terminals are one or more multiples of 360°/n out of phase with the voltages on the other sets. In addition to providing electrical isolation, the transformer may also be selected to step up, step down, or retain the magnitude of the input voltage. One of the phase voltages of each of the n secondary output terminals is applied to a first single pole, n throw switch, a second phase voltage of each of the secondary output terminals is provided to a second single pole, n throw switch, and a third of the output phase voltages from each of the sets of output terminals is applied to a third single pole, n throw switch. The three switches are controlled to switch together so that in each position of the switches the three-phase voltages from one of the sets of the secondary output terminals are connected to the poles of the three switches that in turn are connected to three output terminals of the converter. These three switches may then be switched at a desired frequency and duty cycle to obtain an output voltage at the output terminals of the converter that is at a selected frequency. 
   The multiple sets of three-phase voltages at the sets of secondary output terminals of the transformer form a complete basis set of functions from which any set of three-phase output voltages at any arbitrary frequency and phase angle may be derived by appropriate choice of duty ratio functions. 
   Because the converter of the invention includes a transformer, it is ideally suited for applications where transformer isolation and voltage step up or step down are required, and it provides bidirectional power flow and sinusoidal input and output waveforms. In contrast to conventional DC link conversion systems, a DC link energy storage capacitor is not required, eliminating one of the reliability problem areas of conventional converters, and additional semiconductor switches are not required in contrast to the matrix converter. 
   Further objects, features and advantages of the invention will be apparent from the following detailed description when taken in conjunction with the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings: 
       FIG. 1  is a simplified ideal switch equivalent circuit of a prior art three-phase AC to three-phase AC frequency changer with a DC current link. 
       FIG. 2  is an example of a realization of each pole of the switch for the inverter of  FIG. 1 . 
       FIG. 3  is a simplified ideal switch equivalent circuit of a three-phase AC to three-phase AC frequency changer with a DC voltage link in accordance with the prior art. 
       FIG. 4  is an example of a solid-state switch implementation for each pole of the switch of the inverter of  FIG. 3 . 
       FIG. 5  is an ideal switch equivalent circuit of a three-phase AC to three-phase AC frequency changer with matrix converter topology in accordance with the prior art. 
       FIG. 6  shows a solid state switch realization of each pole of the switch of the matrix converter of  FIG. 5 . 
       FIG. 7  is an ideal switch equivalent circuit schematic of a three-phase AC to three-phase AC frequency converter in accordance with the invention. 
       FIG. 8  is an example of a solid-state switch realization of the switches for the frequency converter of  FIG. 7 . 
       FIG. 9  is a more detailed schematic circuit diagram illustrating the connections of the transformer and the solid state switches for the frequency converter in accordance with the invention. 
       FIG. 10  are phasor representations of three-phase systems of voltages feeding the voltage port of the AC to AC frequency converter in accordance with the invention. 
       FIG. 11  is a simplified schematic of the circuit of  FIG. 9  illustrating the transformer connections and corresponding output voltage phasors to drive the set of orthogonal three-phase systems in accordance with the invention (windings that are parallel to each other in  FIG. 11  are coupled together between the primary and secondary). 
       FIG. 12  are waveforms obtained from a simulation of the AC to AC frequency converter of the invention using vector switching with a 60 Hz input and a 200 Hz output. 
       FIG. 13  are waveforms obtained from a simulation of the AC to AC frequency converter of the invention using vector switching with a 60 Hz input and a 20 Hz output. 
       FIG. 14  is a block diagram of a controller for the frequency converter of the invention. 
       FIG. 15  is a simplified single phase equivalent circuit schematic diagram illustrating bidirectional power flow in the converter of the invention. 
       FIG. 16  is a circuit diagram illustrating the connections of the transformer and the solid-state switches for a frequency converter in accordance with the invention having a transformer with three sets of three-phase secondaries. 
       FIG. 17  is a phasor diagram of the secondary voltages for one phase for the converter of  FIG. 9 . 
       FIG. 18  is a phasor diagram of the secondary voltages for one phase for the converter of  FIG. 16 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   For purposes of illustrating the invention, a frequency converter in accordance with the present invention having four sets of transformer secondaries is shown generally at  40  in schematic form in  FIG. 7 . It is understood that the present invention may be implemented with a transformer having n sets of secondary terminals having voltages which are multiples of 360°/n out of phase, where n is three or more. The converter  40  receives three-phase input power at input terminals  41 , with phase voltages V iA , V iB , V iC , and provides output power at output terminals  42 , with phase voltages V 10A , V 10B , V 10C  and output currents I 10A , I 10B , I 10C . In accordance with the invention, the frequency of the output voltages and currents at the output terminals  42  may be selected to be different from (either higher or lower than) the frequency of the voltage applied to the input terminals  41 . In addition, the magnitude of the output voltages V 10A , V 10B , V 10C , may, if desired, also be higher or lower than the magnitude of the input voltages V iA , V iB , V iC . 
   The power converter  40  includes a transformer  45  which receives the input voltages at the terminals  41  and provides four sets of three-phase output voltages at sets of secondary output terminals  46 ,  47 ,  48  and  49 . The three-phase voltages at each of the sets of output terminals  46 ,  47 ,  48  and  49  are one or more multiples of 90° out of phase with the voltages on the other sets of output terminals. The transformer  45  provides electrical isolation between the input terminals  41  and the secondary output terminals  46 – 49 , and the transformer  45  may be selected to step-up, step-down, or retain the magnitude of the input voltage on the input terminals  41  at the secondary output terminals  46 – 49 . One of the phase voltages of each of the secondary output terminals  46 – 49  is applied to a first single pole, four throw switch  51 , a second phase voltage of each of the secondary output terminals  46 – 49  is applied to a second single pole, four throw switch  52 , and a third of the output phase voltages from each of the sets of output terminals  46 – 49  is applied to a third single pole, four throw switch  53 . Output lines  54 ,  55  and  56  from the poles of the switches  51 ,  52  and  53 , respectively, are connected to the output terminals  42 . The switching of the switches  51 – 53  is actuated by control signals provided from a controller  57 . Each of the switches  51 ,  52  and  53  may be realized utilizing four gate controlled switching devices  58  (e.g., IGBTs with anti-parallel connected diodes) which are connected together at a common node  59  that forms the output pole of the switch  51 ,  52  or  53 , and that may be connected to the output line  54 ,  55  or  56 , as illustrated in  FIG. 8 . 
     FIG. 9  illustrates an example of an implementation of the frequency converter  40  utilizing a transformer  45  having a star connected primary  60 , a parallel, delta connected primary  61 , and four star connected secondaries  64 ,  65 ,  66  and  67 . The secondaries  64  and  65  are coupled to the star connected primary  60  and the secondaries  66  and  67  are coupled to the delta connected primary  61 . Each of the secondary windings is magnetically coupled to the primary winding to which it is shown parallel in  FIG. 9 . For purposes of exemplification, in  FIG. 9  the output voltages at the output terminals  42  are supplied through smoothing inductors  70  to a three-phase motor  71  to drive the motor at a variable frequency that is determined by the switching of the switches  51 – 53 , as explained further below. A generator may be connected in place of the motor  71  since power can be transferred in either direction (or the motor  71  can function as a generator in a regenerative braking mode). Capacitor-inductor filters may also be connected to the lines leading to the motor  71  to provide further filtering of higher frequency noise from the power provided to the motor, and capacitors or snubber circuits may be connected to the secondary output terminals to reduce transient voltage spikes. 
   The theory of operation of the frequency converter  40  and the manner in which the switches  51 – 53  may be controlled to provide a selected output frequency are discussed below. 
   Let the voltages and currents at the voltage port (input terminals) be defined as V iS =[V iSA  V iSB  V iSC ] T , and I iS =[I iSA  I iSB  I iSC ] T , respectively for i=1 . . . 4. Let the voltages and currents at the current port (output terminals) be defined as V 10 =[V 10A  V 10B  V 10C ] T , I 10 =[I 10A  I 10B  I 10C ] T , respectively. Let H ij (t), the switching function of the throws of the switches, be defined as 
                 H     i   ⁢           ⁢   10       ⁡     (   t   )       =     {               1         if   ⁢           ⁢     t     i   ⁢           ⁢   10       ⁢             ⁢             ⁢   is   ⁢           ⁢   closed             0       otherwise         ⁢           ⁢   for   ⁢           ⁢   i     =   1     ,     2   ⁢           ⁢   …   ⁢           ⁢   4                 (   1   )             
 
   Note that all the throws of the three poles of the switch are “ganged” together so that they operate in synchronism. The transfer properties of the converter may now be defined as 
                 V   10     ⁡     (   t   )       =       ∑     i   =   1     4     ⁢           ⁢         H     i   ⁢           ⁢   10       ⁡     (   t   )       ·     V   iS                 (   2   )             
 
and
 
 I   is ( t )= H   i10 ( t )· I   10   (3)
 
   The average value of the switching functions may be readily represented by the duty ratio of the particular throw using 
                 m     i   ⁢           ⁢   10       ⁡     (   τ   )       =       1   T     ⁢       ∫     τ   -   T     τ     ⁢         H     i   ⁢           ⁢   10       ⁡     (   t   )       ·           ⁢     ⅆ   t                   (   4   )             
 
   With the definition of the average switching function (or duty ratio of the i th  throw), the transfer properties now become, 
                 V   10     ⁡     (   t   )       =       ∑     i   =   1     4     ⁢           ⁢         m     i   ⁢           ⁢   10       ⁡     (   t   )       ·     V   iS                 (   5   )             
 
and
 
 I   is ( t )= m   i10 ( t )· I   10   (6)
 
   Let the set of three-phase voltages be chosen as
 
 V   1S   =V   m [cos(θ)cos(θ−2π/3)cos(θ+2π/3)] T ,  (7)
 
 V   2S   =−V   m [cos(θ)cos(θ−2π/3)cos(θ+2π/3)] T ,  (8)
 
 V   3S   =V   m [sin(θ)sin(θ−2π/3)sin(θ+2π/3)] T ,  (9)
 
 V   4S   =−V   m [sin(θ)sin(θ−2π/3)sin(θ+2π/3)] T ,  (10)
 
where V m  is the peak value of the input line to neutral voltage and θ=2π Ft, F being the input frequency.  FIG. 10  represents the phasors of the four three-phase systems of voltages. These four sets of three-phase voltages form an orthogonal set of functions from which any set of three-phase output voltages may be derived through appropriate choice of duty ratio functions. For instance, let the duty ratio function for the different throws be chosen as
 
 m   110 ( t )=[1+cos(β+θ)]/4 ;m   210 ( t )= m [1−cos(β+θ)]/4  (10)
 
 m   310 ( t )= m [1+sin(β+θ)]/4 ; m   410 ( t )= m [1−sin(β+θ)]/4;  (11)
 
where m is a selected modulation index and β=2πF o t, with F o  being the desired output frequency. With this choice of duty ratio functions the output voltages become,
 
 V   10   =mV   m [cos(β)cos(β−2π/3)cos(β+2π/3)] T 
 
thus realizing the frequency conversion function. In order to derive the orthogonal set of three-phase voltages, the transformer  45 , comprised of four three-phase transformers, may be used, as illustrated in  FIG. 11  in simplified form. Duty ratio functions other than those set forth in equations (10) and (11) may also be utilized.
 
   Detailed computer simulation of the converter operation and modulation algorithm in accordance with the invention was carried out and the results are illustrated in  FIG. 12 . The simulation model included a three-phase L-C output filter that is not illustrated in  FIG. 9 . The waveforms shown in  FIG. 12  are the three-phase input voltages (top), output currents (middle), and output voltages (bottom). The operating conditions are input voltage at 60 Hz, and output voltage at 200 Hz.  FIG. 13  shows corresponding waveforms when the frequency is changed from 60 Hz input to 20 Hz output. 
   A block diagram of the controller  57  that may be utilized to carry out control of the switches of the converter is shown in  FIG. 14 . The controller  57  illustratively includes a digital controller  72  such as a digital signal processor (DSP) of conventional implementation for control of AC machine drives, e.g., Texas Instruments TMS 320F240 and Motorola 56F801. The digital controller  72  receives signals corresponding to measured input voltages V iA , V iB , V iC  (from the input terminals  41 ) and output currents I OA , I OB , I OC  (at the output terminals  42 ) on lines  73 , which signals are converted to digital data by an analog to digital converter  74 . Command values for voltage, currents, and power throughput are received from a user interface  75 . The digital controller  57  performs duty ratio calculations as discussed above and protection functions, and provides output signals through a digital output interface  76  to gate drivers  77  that provide the gate drive signals to the switches (e.g., IGBTs) of the converter. 
   The converter of the invention can also carry out control of power flow without changing frequency. This may be illustrated with respect to the diagram of  FIG. 15 , a single-phase equivalent circuit of the three-phase system, in which the converter  40  interconnects a sending voltage source  80  of voltage V S  and a receiving voltage source  81  of voltage V R , with a line reactance  83  having a reactance value X. In this case, if the duty ratio of the various throws are as described above, the complex representation of the power received may be expressed as: 
             S   =             [       (       m   110     -     m   210       )     +     j   ⁡     (       m   410     -     m   310       )         ]     ⁢     V   S       -     V   R         j   ⁢           ⁢   X       ⁢           ⁢     V   R   *               (   12   )             
 
where the voltages are represented as complex phasors and * represents the complex conjugate phasor.
 
   If the vector switching converter  40  was not present in the system the power received is: 
       S   =           V   S     -     V   R         j   ⁢           ⁢   X       ⁢           ⁢     V   R   *           
 
From equations (12) and (13), it may be noted that by suitably modifying the duty ratio variables, the real and reactive power transferred through the line may be controlled appropriately. In some cases, it may be desirable to operate the switching power converter  40  in conjunction with conventional power flow control devices such as mechanical or thyristorized tap changing transformers and boosters in order to improve the controllability at an economical cost.
 
   As discussed above, the present invention may be implemented with a transformer having three or more sets of secondary terminals.  FIG. 16  is a circuit diagram of an implementation of the invention with a transformer having a star connected three-phase primary  90  and three star connected three-phase secondaries  91 – 93  which are connected to three sets of output terminals  95 – 97 , with each of the switching devices  58  within each of the switches  51 – 53  being connected to one of the secondary output terminals. Each of the secondary windings is magnetically coupled to the primary winding to which it is shown parallel in  FIG. 16 . The implementation of  FIG. 16  requires one less secondary and one less switching device  58  in each of the switches  51 – 53  than the implementation of  FIG. 9 , but with a corresponding reduction in realizable output voltage magnitude. For a converter of the invention having “n” sets of secondary windings, the largest amplitude of any phase of the balanced three-phase output voltages is limited by the radius of the largest circle that can be drawn inside the polygon formed by the phasors of the secondary voltages V 1A , V 2A , . . . V nA , V 1B , V 2B , . . . V nB , and V 1C , V 2C , . . . V nC . The circle and polygon formed by the phasors is shown in  FIG. 17  for four sets of secondary windings and is shown in  FIG. 18  for three sets of secondary windings. These figures illustrate the reduction in available output voltage that occurs with a reduction in secondary terminals. If desired, the secondary windings can be wound with a larger number of turns to compensate for the reduction in voltage. Appropriate tradeoffs in the design of the converter can be made to fulfill design objectives for a specific application. 
   The number of converter semiconductor switching devices (e.g., IGBTs) utilized in the converter of the invention compares favorably with matrix converters (12 for a converter having four secondaries rather than 18 required for a comparable matrix converter) and equally with the DC link converter approaches. Since the converter of the invention utilizes a transformer, it is ideally suitable for applications where transformer isolation and voltage step-down or step-up are required, and is capable of providing bidirectional power flow and sinusoidal input and output waveforms. 
   It is understood that the invention is not confined to the particular embodiments set forth herein, but embraces all such forms thereof as come within the scope of the following claims.