Patent Publication Number: US-8111835-B2

Title: Active noise control apparatus

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to an active noise control apparatus for reducing an in-compartment noise with a cancellation sound, which is opposite in phase to the in-compartment noise, and more particularly to an active noise control apparatus for reducing a drumming noise (hereinafter also referred to as “road noise”), which is generated in the passenger compartment of a vehicle while the vehicle is running. 
     2. Description of the Related Art 
     Heretofore, there has been known in the art an active noise control apparatus (hereinafter also referred to as a “periodic-noise-compatible ANC”) for reducing noise (hereinafter referred to as “engine muffled sound” or “engine noise”) caused by a vibratory noise, which is produced by a vibratory noise source such as an engine or the like on a vehicle and generated periodically in synchronism with the rotation of the engine, by generating a control signal via a control unit, for canceling the engine noise based on a signal that is highly correlated to the vibratory noise produced by the vibratory noise source, and outputting a canceling sound, which is opposite in phase to the engine noise based on the control signal, into the passenger compartment of the vehicle (see Japanese Laid-Open Patent Publication No. 2004-361721). 
     While the vehicle is running, vibrations of the tires caused by the road are transmitted through suspensions to the vehicle body, thereby producing an aperiodic drumming noise (road noise) in the passenger compartment. The road noise constitutes a non-periodically generated low-frequency noise generated in the passenger compartment, and is produced as a resonant sound having a high sound pressure level at a certain frequency (resonant frequency), due to the resonant characteristics of the passenger compartment. Therefore, the resonant sound is defined by the road noise having a central frequency equal to a certain resonant frequency f of 40 [Hz], for example. 
     Japanese Laid-Open Patent Publication No. 2000-322066 discloses an active noise control apparatus (hereinafter also referred to as an “aperiodic-noise-compatible ANC”) including a plurality of microphones installed in a passenger compartment. The microphones generate canceling error signals based on differences (hereinafter also referred to as “canceling error sounds”) between the noise in the passenger compartment and a canceling sound, and output the generated canceling error signals to a control unit. The control unit generates a control signal based on the canceling error signals, and a speaker outputs a canceling sound based on the control signal into the passenger compartment. In this manner, road noise is reduced by the canceling sound according to a feedforward control process. Japanese Laid-Open Patent Publication No. 2000-322066 also reveals that microphones are used to detect noise in the passenger compartment, a control unit, which is in the form of an analog circuit, generates a control signal based on the noise, and that a speaker outputs canceling sounds based on the control signal into the passenger compartment. In this manner, road noise is reduced by canceling sounds according to a feedback control process. 
     Japanese Laid-Open Patent Publication No. 2001-282255 discloses that a speaker is shared by a periodic-noise-compatible ANC and/or an aperiodic-noise-compatible ANC (hereinafter also referred to simply as an “ANC”) and an audio system on a vehicle, so that the speaker can output sounds based on an output signal from the audio system, and a canceling sound based on a control signal from the ANC, into the passenger compartment of the vehicle. 
     Engine noise referred to above is defined as periodically generated noise within a narrow frequency band having a predetermined central frequency. A periodic-noise-compatible ANC generates a control signal having a control frequency depending on the predetermined central frequency, and the speaker outputs canceling sounds having the control frequency into the passenger compartment for effectively reducing noise in the passenger compartment. 
     Road noise is defined as aperiodically generated low-frequency noise having a central frequency equal to a resonant frequency of 40 [Hz], for example, which is determined from the resonant characteristics of the passenger compartment. An aperiodic-noise-compatible ANC is required to reduce resonant sounds at respective resonant frequencies. 
     If the aperiodic-noise-compatible ANC generates a control signal according to a feedforward control process, then the control unit needs to comprise a FIR adaptive filter and a DSP (Digital Signal Processor) for performing convolutional calculations at the respective resonant frequencies. As a result, the aperiodic-noise-compatible ANC is relatively expensive to manufacture. Furthermore, since the aperiodic-noise-compatible ANC generates a control signal at the resonant frequencies, while updating the filter coefficient of the adaptive filter from time to time, the control unit suffers from an increased computational burden in connection with generating the control signal. 
     If the aperiodic-noise-compatible ANC generates a control signal according to a feedback control process, then the control unit needs to comprise a combination of several analog filters for generating a control signal at the resonant frequencies. As a result, the control unit requires a large circuit scale, thereby causing the ANC including the control unit to have a large unit size. However, it is difficult for an ANC having such a large unit size to find sufficient installation space inside the vehicle. In addition, it is also difficult to combine the ANC having such a large unit size with a digital audio unit. 
     An aperiodic-noise-compatible ANC has been considered for generating a control signal according to a feedback control process based on a digital signal processing method, to thereby silence an aperiodic resonant sound (resonant noise). 
       FIG. 18  of the accompanying drawings shows an aperiodic-noise-compatible ANC  200  comprising a microphone (canceling error signal detector)  18  and a speaker (sound output device)  22 , which are disposed in the passenger compartment  14  of a vehicle, and a control unit  50 . The control unit  50  comprises an A/D converter (ADC)  59 , a controller  202  in the form of a microcomputer and having a given transfer function H, and a D/A converter (DAC)  65 . The aperiodic noise includes a resonant sound (aperiodic resonant noise), which is aperiodically generated inside the passenger compartment  14 , and which has a high sound pressure level at a certain resonant frequency f due to the configuration of the passenger compartment  14 . 
     It is assumed that, at a time t(n−1) of a sampling event (n−1), the controller  202  generates a control signal y(n−1) in the form of a digital signal for canceling out noise (aperiodic noise) in the passenger compartment  14 . Then, the DAC  65  converts the control signal y(n−1) into an analog signal, and the speaker  22  outputs a canceling sound into the passenger compartment  14  for canceling out the noise, based on the analog control signal y(n−1). 
     The microphone  18  is located at an antinode of the acoustic mode of the passenger compartment  14 . At a time t(n) of a sampling event n, the microphone  18  outputs a canceling error signal e(n) to the ADC  59 , representing a difference (canceling error sound) between the canceling sound and the noise. 
     Specifically, at the sampling event n, a canceling sound at the position of the microphone  18  is defined as a canceling sound that has been output from the speaker  22 , based on the control signal y(n−1) from the controller  202  at the preceding sampling event (n−1), and that has reached the microphone  18 . If the transfer characteristics from the speaker  22  to the microphone  18  with respect to the sound at the resonant frequency f are represented by C, then the canceling sound (the signal depending thereon) at the position of the microphone  18  at the sampling event n is represented by C·y(n−1). The transfer characteristics C are divided into gain characteristics (amplitude change) G′ and a phase delay (phase characteristics) φ′. At the sampling event n, the resonant noise (the signal depending thereon) having a resonant frequency f at the position of the microphone  18  is represented by d(n). 
     Therefore, the canceling error signal e(n) output from the microphone  18  to the ADC  59  is expressed according to the following equation (1):
 
 e ( n )= d ( n )+ C·y ( n− 1)  (1)
 
     The ADC  59  converts the canceling error signal e(n) from an analog signal into a digital signal, and outputs the digital canceling error signal e(n) as an input signal x(n) to the controller  202 . Based on the input signal x(n) {=e(n)}, the controller  202  generates a control signal y(n) {=−d(n+1)/C} depending on the canceling sound C·y(n), which is opposite in phase with a resonant noise d(n+1) at the position of the microphone  18 . 
     According to the silencing control process carried out by the ANC  200  to silence the resonant noise, it is important to decide how to generate the control signal y(n) for the canceling sound C·y(n), which is opposite in phase with the resonant noise d(n+1) at the position of the microphone  18 . 
     If it is assumed that the control signal n(y−1) is generated at the preceding sampling event (n−1) and the resonant noise d(n) at the position of the microphone  18  happens to be completely silenced by the canceling sound C·y(n−1) at the present sampling event n, then since the canceling error signal e(n) output from the microphone  18  is expressed by e(n)=d(n)+C·y(n−1)=0, the signal x(n) input to the controller  202  is expressed by x(n)=e(n)=0. 
     Since x(n)=0 regardless of the resonant noise d(n) present at the sampling event n, the controller  202  is unable to generate a control signal y(n) and the speaker  22  is unable to output a canceling sound. Therefore, the resonant noise d(n+1) at the position of the microphone  18  cannot be silenced. Alternatively, the controller  202  fails to generate a highly accurate control signal y(n), and even if the speaker  22  outputs a canceling sound, the resonant noise d(n+1) at the position of the microphone  18  cannot be silenced completely and the resonant noise d(n+1) remains unsilenced. As a result, the resonant noise d(n+1) cannot be silenced stably at the next sampling event (n+1). 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide an active noise control apparatus, which is capable of generating a control signal according to a simple digital signal processing method, benefits from a reduced computational burden in generating the control signal, and is relatively inexpensive to manufacture. 
     Another object of the present invention is to provide an active noise control apparatus, which is capable of stably silencing road noise in order to reliably reduce the road noise. 
     For easier understanding of the present invention, various elements and items shall be described below in combination with reference numerals and characters used in the accompanying drawings. However, such elements and items should not be interpreted as being limited to the components, signals, and other properties that are accompanied by these reference numerals and characters. 
     An active noise control apparatus (ANC)  204  according to the present invention basically comprises a control unit  50  for generating a control signal y(n), y(n−1) for canceling out noise in a passenger compartment  14  of a vehicle  12 , a sound output device  22  for outputting a canceling sound for canceling out the noise based on the control signal y(n), y(n−1) into the passenger compartment  14 , and a canceling error signal detector  18  for outputting a canceling error signal e(n) representing a canceling error sound between the noise and the canceling sound to the control unit  50 . 
     According to a first aspect of the ANC  204 , as shown in  FIGS. 1 through 5 , the control unit  50  comprises an A/D converter  59  for converting the canceling error signal e(n) from an analog signal into a digital signal, an echo canceler  58  for correcting the control signal y(n−1) and thereby generating a digital echo canceling signal Ĉ·y(n−1) based on a corrective value Ĉ corresponding to (identifying) transfer characteristics between the sound output device  22  and the canceling error signal detector  18 , a subtractor  60  for generating a first basic signal x(n) by subtracting the digital canceling error signal Ĉ·y(n−1) from the digital echo canceling signal e(n), a delay filter  54  for generating a second basic signal x′(n) by delaying the first basic signal x(n) by a time Z −n  corresponding to a ¼ period of a resonant frequency f determined by resonant characteristics of the passenger compartment  14 , a first filter  62  for correcting the first basic signal x(n) and thereby generating a first corrective signal A·x(n), a second filter  64  for correcting the second basic signal x′(n) and thereby generating a second corrective signal B·x′(n), an adder  56  for generating the control signal y(n) by combining the first corrective signal A·x(n) and the second corrective signal B·x′(n), and a D/A converter  65  for converting the control signal y(n) from a digital signal into an analog signal and outputting the analog control signal to the sound output device  22 . 
     The resonant frequency f of a resonant sound, such as road noise, is a known frequency determined by the structure of the passenger compartment. It is desirable for the ANC  204  to be able to reduce the resonant sound (first noise) at the known resonant frequency f. The control unit  50  generates the control signal y(n), which has a control frequency equal to the resonant frequency f and which is in opposite phase with the resonant sound. The sound output device  22  outputs the canceling sound based on the control signal y(n). 
     According to the first aspect, the control unit  50  has the echo canceler  58 , which stores the corrective value Ĉ identifying the transfer characteristics C from the sound output device  22  to the canceling error signal detector  18 , with respect to the sound at the control frequency f. The subtractor  60  subtracts the digital echo canceling signal Ĉ·y(n−1) produced by correcting the control signal y(n−1) with the corrective value Ĉ from the canceling error signal e(n) output from the canceling error signal detector  18 , thereby estimating a noise d(n) to be silenced at the position of the canceling error signal detector  18 . The estimated noise d(n) is represented by the first basic signal x(n) that is supplied to a controller  202 . 
     In the ANC  204 , the first basic signal x(n) is expressed according to the following equation (2):
 
 x ( n )= e ( n )− Ĉ·y ( n− 1)≈ d ( n )  (2)
 
     The corrective value Ĉ corresponding to (identifying) the transfer characteristics C represents signal transfer characteristics from an output terminal of the D/A converter  65  to an output terminal of the A/D converter  59 , including the transfer characteristics C from the sound output device  22  to the canceling error signal detector  18 . 
     The signal transfer characteristics are actually measured as follows: As shown in  FIG. 2 , a signal transfer characteristics measuring device  300 , which comprises a Fourier transforming device, is connected between the input and output terminals of the controller  202 . The signal transfer characteristics measuring device  300  measures signal transfer characteristics based on a test signal input from the controller  202  to the D/A converter  65  and a signal output from the subtractor  60  to the controller  202 . The signal transfer characteristics measured by the signal transfer characteristics measuring device  300  are set as the corrective value Ĉ in the echo canceler  58 . Depending on how the signal transfer characteristics measuring device  300  measures the signal transfer characteristics, the corrective value Ĉ may represent signal transfer characteristics from the sound output device  22  to the canceling error signal detector  18 , or signal transfer characteristics from the output to input terminals of the controller  202 , including the signal transfer characteristics from the sound output device  22  to the canceling error signal detector  18 . 
     The corrective value (transfer characteristics) Ĉ including the transfer characteristics C is identified according to the above measuring process. As with the transfer characteristics C, the transfer characteristics Ĉ are also divided into gain characteristics (amplitude change) G and a phase delay (phase characteristics) φ. 
     In the controller  202 , the delay filter  54  generates the second basic signal x′(n) by delaying the first basic signal x(n) a predetermined time Z −n  based on the control frequency f, and the adder  56  combines a first corrective signal A·x(n) produced by correcting the first basic signal x(n) and a second corrective signal B·x′(n) produced by correcting the second basic signal x′(n), resulting in the control signal y(n). 
     Since the controller  202  generates the control signal y(n) {=−d(n+1)/Ĉ}, for canceling out the noise d(n+1) to be silenced at the position of the canceling error signal detector  18 , from the first basic signal x(n) and the second basic signal x′(n) and based on the noise d(n) estimated by the subtractor  60 , the canceling sound for canceling out the noise can simply and accurately be generated without the need for a FIR adaptive filter. Thus, the ANC  204  has a simpler arrangement and can be manufactured less expensively. 
     Since the first basic signal x(n) is used to represent the noise d(n) that is determined by subtracting the echo canceling signal Ĉ·y(n−1) from the canceling error signal e(n), the control signal y(n) can be generated as long as the noise d(n) is present, so that the noise d(n+1) at the position of the microphone  18  can stably be silenced. 
     If it is assumed that the noise d(n) is not estimated using the canceling signal Ĉ·y(n−1), but the canceling error signal e(n) is directly used as the first basic signal x(n) (see  FIG. 18 ), and a noise d(i) at the position of the canceling error signal detector  18  happens to be completely silenced at a certain instant (sampling event: n=1), then since e(i)=x(i)=0, the controller  202  is unable to generate a control signal y(n) {y(i)=0}, regardless of the noise d(i) being present in the passenger compartment  14 , and the speaker  22  is unable to output any canceling sounds. Therefore, the noise d(n+1) at the position of the canceling error signal detector  18  cannot be silenced in the next sampling event (n=i+1). Alternatively, the controller  202  fails to generate a highly accurate control signal y(i), and even if the speaker  22  outputs a canceling sound, the noise d(n+1) at the position of the canceling error signal detector  18  cannot be silenced completely, but rather remains unsilenced. As a result, the noise d(n+1) cannot stably be silenced. 
     The predetermined time Z −n  corresponds to π/2 (90°}, and the first basic signal x(n) and the second basic signal x′(n) are orthogonal to each other on a Gaussian plane, as shown in  FIG. 3C . 
     According to a second aspect of the ANC  204 , the control unit  50  comprises an A/D converter  59  for converting the canceling error signal e(n) from an analog signal into a digital signal, an echo canceler  58  for correcting the control signal y(n−1) and thereby generating a digital echo canceling signal Ĉ·y(n−1) based on transfer characteristics (a corrective value Ĉ identifying such transfer characteristics) between the sound output device  22  and the canceling error signal detector  18 , a subtractor  60  for generating a first basic signal x(n) by subtracting the digital canceling error signal Ĉ·y(n−1) from the digital echo canceling signal e(n), a delay filter  55  for generating a second basic signal x″(n) by delaying the first basic signal x(n) by a predetermined time Z −m  based on a resonant frequency f determined by resonant characteristics of the passenger compartment  14 , an adder  56  for combining the first basic signal x(n) and the second basic signal x″(n) into a combined signal {x(n)+x″(n)}, an amplitude adjuster  70  for adjusting the amplitude of the combined signal {x(n)+x″(n)} with a predetermined gain P to a predetermined magnitude thereby generating the control signal y(n), and a D/A converter  65  for converting the control signal y(n) from a digital signal into an analog signal and outputting the analog control signal to the sound output device  22 . 
     The first aspect is different from the second aspect as to the predetermined time Z −m  by which the first basic signal x(n) is delayed. As with the transfer characteristics Ĉ in the first aspect, the transfer characteristics (the corrective value) Ĉ is divided into gain characteristics (amplitude change) G and a phase delay (phase characteristics) φ. 
     Specifically, the predetermined time Z −m  has a value based on the control frequency f and phase characteristics (phase delay φ) of the transfer characteristics Ĉ with respect to the sound at the control frequency f. Specifically, the predetermined time Z −m  is a time corresponding to a phase value 2Ψ that is twice the value produced by subtracting the phase characteristics (phase delay) φ from the phase difference between the first basic signal x(n) and the canceling sound Ĉ·y(n), which is opposite in phase with the noise d(n). 
     The predetermined time Z −m  actually is determined on a trial and error basis, based on the gain P of the amplitude adjuster  70  and a phase value Ψ, at the time a test noise d(n) having the control frequency f is generated in the passenger compartment  14  and the generated test noise is silenced at the position of the canceling error signal detector  18 . 
     According to the first and second aspects, since the control signal y(n) is simply and accurately generated without the need for a conventional FIR adaptive filter, and is output as a canceling sound from the sound output device  22  into the passenger compartment  14 , drumming noises including road noises in the passenger compartment  14  can reliably be reduced. 
     Particularly, according to the first aspect, inasmuch as the second basic signal x′(n) is generated by delaying the first basic signal x(n) by a time Z −n  corresponding to a ¼ period of the resonant frequency (control frequency) f, that is, by shifting the phase of the first basic signal x(n) by 90°, the control signal y(n) {=−d(n+1)/Ĉ} for canceling out the noise d(n+1) to be silenced at the position of the canceling error signal detector  18  can simply and accurately be generated from the first basic signal x(n) and the second basic signal x′(n). Thus, the ANC  204  has a simpler arrangement and can be manufactured less expensively. 
     Since the control unit  50  can generate the control signal y(n) through a simpler digital signal processing method, the computational burden for generating the control signal y(n) is reduced. Further, since the control unit  50  may be of a simple arrangement using a microcomputer  52 , which is relatively inexpensive, the ANC  204  can be manufactured inexpensively. As a result, the ANC  204  can be reduced in overall size, and the ANC  204  be combined with a digital audio unit in the vehicle  12 . 
     According to the first aspect, the echo canceler  58  preferably comprises a first cosine corrector  80  for correcting the first basic signal x(n) with a cosine value Cr of phase characteristics (phase delay φ) of the transfer characteristics Ĉ and outputting a corrected signal, a first sine corrector  82  for correcting the second basic signal x′(n) with a sine value Ci of the phase characteristics and outputting a corrected signal, a subtractor  88  for subtracting the corrected signal output from the first sine corrector  82  from the corrected signal output from the first cosine corrector  80  thereby to generate a differential signal Sm, a second cosine corrector  84  for correcting the second basic signal x′(n) with the cosine value Cr and outputting a corrected signal, a second sine corrector  86  for correcting the first basic signal x(n) with the sine value Ci and outputting a corrected signal, a first adder  90  for adding the corrected signal output from the second cosine corrector  84  and the corrected signal output from the second sine corrector  86  into a sum signal Sp, a first correcting filter  92  for correcting the differential signal Sm and outputting a corrected signal, a second correcting filter  94  for correcting the sum signal Sp and outputting a corrected signal, and a second adder  96  for adding the corrected signal from the first correcting filter  92  and the corrected signal from the second correcting filter  94  together with the echo canceling signal Ĉ·y(n−1), and outputting the echo canceling signal Ĉ·y(n−1) to the subtractor  60 . 
     The processing sequence for generating the echo canceling signal Ĉ·y(n−1) in the echo canceler  58  comprises a total of nine processes including arithmetic operations, i.e., four correcting processes carried out respectively by the first cosine corrector  80 , the second cosine corrector  84 , the first sine corrector  82 , and the second sine corrector  86 , one subtracting process carried out by the subtractor  88 , one adding process carried out by the first adder  90 , two correcting processes carried out respectively by the first correcting filter  92  and the second correcting filter  94 , and one adding process carried out by the second adder  96 . As a result, the amount of processing operations required for generating the echo canceling signal is reduced. In other words, the echo canceling signals Ĉ·y(n−1), Ĉ·y(n) can be generated by a simple arrangement, without the need for a FIR filter. 
     Each of the first filter  62 , the second filter  64 , the first correcting filter  92 , and the second correcting filter  94  should preferably comprise an adaptive filter. The control unit  50  should preferably further comprise a first filter coefficient updater  100  for updating respective filter coefficients A of the first filter  62  and the first correcting filter  92  in order to minimize the canceling error signal e(n) based on the canceling error signal e(n) and the differential signal Sm, a second filter coefficient updater  102  for updating respective filter coefficients B of the second filter  64  and the second correcting filter  94  in order to minimize the canceling error signal e(n) based on the canceling error signal e(n) and the sum signal Sp. 
     Accordingly, even if the transfer characteristics C, Ĉ vary due to mass-production-induced variations in the layout of the sound output device  22  and the canceling error signal detector  18  in the passenger compartment  14 , or change due to aging or the like, since the filter coefficients A of the first filter  62  and the first correcting filter  92  and the filter coefficients B of the second filter  64  and the second correcting filter  94  are updated under an adaptive control, noise inside the passenger compartment  14  can accurately be silenced. 
     If each of the first filter  62 , the second filter  64 , the first correcting filter  92 , and the second correcting filter  94  comprises an adaptive notch filter, then road noise at a frequency f can reliably be silenced. 
     The control unit ( 50 ) preferably further comprises a delay filter D/A converter  75  and a delay filter A/D converter  77 . The delay filter  74  preferably comprises an allpass filter for equalizing the phase delay at the control frequency f to a phase delay corresponding to a ¼ period of the control frequency f. The delay filter D/A converter  75  preferably should convert the first basic signal x(n) from a digital signal into an analog signal, for outputting the analog first basic signal x(n) to the delay filter  74 . The delay filter A/D converter  77  preferably should convert the second basic signal x′(n) from an analog signal into a digital signal, for outputting the digital second basic signal x′(n) to the second filter  64 . 
     Thus, the delay filter  74  may be in the form of an analog circuit. If the control unit  50  is implemented by a microcomputer, the delay filter  74  needn&#39;t be included in the microcomputer, and hence the microcomputer may be of a simpler arrangement. 
     The ANC  204  preferably comprises an antialiasing filter  66  for passing only a signal having a predetermined frequency or lower, and outputting the signal to the control unit  50 . The predetermined frequency preferably should be higher than a control frequency of the control signal. 
     If the control unit  50  includes a microcomputer for generating the control signal y(n) according to a digital signal processing method, then the antialiasing filter  66  removes a folding noise having a predetermined frequency or higher from the canceling error signal e(n), and then supplies the canceling error signal e(n) to the microcomputer. Accordingly, the control signal y(n) can be generated accurately in the microcomputer. 
     The ANC  204  preferably further comprises a reconstruction filter  68  for removing a high-frequency component from the control signal y(n) output from the control unit  50  and for outputting the control signal y(n) from which the high-frequency component has been removed to the sound output device  22 . The high-frequency component preferably has a frequency higher than the control frequency f. 
     If the control unit  50  includes a microcomputer for generating the control signal y(n) according to a digital signal processing method, and the control signal y(n) is converted into an analog signal that is output to the sound output device  22 , then the reconstruction filter  68  removes a high-frequency component from the analog control signal y(n), so that the analog control signal y(n) exhibits a smooth waveform over time. As a result, the sound output device  22  can output a canceling sound of high quality, based on the control signal y(n) from which the high-frequency component has been removed. 
     The ANC  204  preferably further comprises a bandpass filter  72  for passing and outputting to the control unit  50 , from within the canceling error signal e(n), only a signal in a predetermined frequency band and having a central frequency equal to the control frequency f. 
     If the control unit  50  includes a microcomputer for generating the control signal y(n) according to a digital signal processing method, then the bandpass filter  72  passes only a signal having a predetermined frequency band of the canceling error signal e(n), and the signal that has passed through the bandpass filter  72  is supplied to the microcomputer. Accordingly, the control signal y(n) can be generated accurately in the microcomputer. 
     The above and other objects, features, and advantages of the present invention will become more apparent from the following description when taken in conjunction with the accompanying drawings, in which preferred embodiments of the present invention are shown by way of illustrative example. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic block diagram of an ANC, which is illustrative of a first fundamental concept of the present invention; 
         FIG. 2  is a schematic block diagram illustrating measurement of signal transfer characteristics by a signal transfer characteristics measuring device in the ANC shown in  FIG. 1 ; 
         FIG. 3A  is a diagram of vectors on a Gaussian plane, showing the relationship between Ĉ·y(n) and d(n+1); 
         FIG. 3B  is a diagram of vectors on a Gaussian plane, showing the relationship between Ĉ·y(n) and G-y(n); 
         FIG. 3C  is a diagram of vectors on a Gaussian plane, showing the generation of y(n) from x(n) and x′(n); 
         FIG. 4  is a diagram illustrating the generation of a second basic signal in the case that a delay filter shown in  FIG. 1  comprises buffers; 
         FIG. 5  is a diagram illustrating the generation of a second basic signal in the case that a delay filter shown in  FIG. 1  comprises registers; 
         FIG. 6  is a schematic block diagram of an ANC, which is illustrative of a second fundamental concept of the present invention; 
         FIG. 7  is a diagram of vectors on a Gaussian plane, showing the generation of y(n) from x(n) and x″(n); 
         FIG. 8  is a schematic block diagram of an arrangement of an ANC according to a first embodiment of the present invention; 
         FIG. 9  is a schematic block diagram of an internal arrangement of an ANC electronic controller shown in  FIG. 8 ; 
         FIG. 10  is a schematic block diagram of an arrangement of an ANC according to a second embodiment of the present invention; 
         FIG. 11  is a schematic block diagram of an arrangement of an ANC according to a third embodiment of the present invention; 
         FIG. 12  is a characteristic diagram showing sound pressure vs. frequency characteristics of noise at the position of a microphone; 
         FIG. 13  is a schematic block diagram of an arrangement of an ANC according to a fourth embodiment of the present invention; 
         FIG. 14  is a schematic block diagram of an arrangement of an ANC according to a fifth embodiment of the present invention; 
         FIG. 15  is a schematic block diagram of an arrangement of an ANC according to a sixth embodiment of the present invention; 
         FIG. 16  is a schematic block diagram of an arrangement of an ANC according to a seventh embodiment of the present invention; 
         FIG. 17  is a schematic block diagram of an arrangement of an ANC according to an eighth embodiment of the present invention; and 
         FIG. 18  is a schematic block diagram of an arrangement of an aperiodic-noise-compatible ANC, according to the related art. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Like or corresponding parts are denoted by like or corresponding reference characters throughout the views. 
     Active noise control apparatuses according to preferred embodiments of the present invention shall be described below with reference to the drawings. Prior to describing the specific details of such active noise control apparatuses (hereinafter also referred to as “ANCs”) according to preferred embodiments of the present invention, fundamental concepts (first and second concepts) thereof will be described below with reference to  FIGS. 1 through 7 . 
     With respect to the first and second concepts, parts thereof which are identical to those of the ANC  200  (see  FIG. 18 ) according to the related art shall be denoted by identical reference characters. 
       FIG. 1  is a schematic block diagram of an ANC  204 , to which a first fundamental concept of the present invention is applied. 
     As shown in  FIG. 1 , the ANC  204  is an aperiodic-noise-compatible ANC based on a feedforward control process. The ANC  204  comprises a microphone (canceling error signal detector)  18  and a speaker (sound output device)  22  which are disposed in a passenger compartment  14  of a vehicle, and a control unit  50 . The control unit  50  comprises an ADC (A/D converter)  59 , an echo canceler  58 , a subtractor  60 , a controller  202 , and a DAC (D/A converter)  65 .  FIG. 1  illustrates operation of the ANC  204  in a sampling event n at a given time t(n). Similarly, operation of the ANCs shown in other block diagrams will also be described as occurring in a sampling event n at a given time t(n). 
     The controller  202  includes a transfer function H, and comprises a first filter  62  having a filter coefficient (gain) A, a second filter  64  having a filter coefficient (gain) B, a delay filter  54 , and an adder  56 . 
     It is assumed that at time t(n−1) of a sampling event (n−1), the controller  202  generates a control signal y(n−1) in the form of a digital signal for canceling out noise in the passenger compartment  14 . The DAC  65  converts the control signal y(n−1) into an analog signal, and the speaker  22  outputs a canceling sound into the passenger compartment  14  for canceling out noise based on the analog third control signal y(n−1). 
     The microphone  18  is located at an antinode of the acoustic mode of the passenger compartment  14 . At a given sampling event n, the microphone  18  outputs a canceling error signal e(n), representing the difference (canceling error sound) between the canceling sound and the noise to the ADC  59 . The noise includes a resonant sound (aperiodic resonant noise), which is aperiodically generated in the passenger compartment  14  and has a high sound pressure level at a certain resonant frequency f, due to the configuration of the passenger compartment  14 . 
     The ADC  59  converts the canceling error signal e(n) from an analog signal into a digital signal, and outputs the digital canceling error signal e(n) to the subtractor  60 . The echo canceler  58  generates an echo canceling signal Ĉ·y(n−1) by correcting the control signal y(n−1) with a corrective value Ĉ, which is representative of transfer characteristics C from the speaker  22  to the microphone  18  with respect to the sound of a control frequency f. Then, the echo canceler  58  outputs the generated echo canceling signal Ĉ·y(n−1) to the subtractor  60 . The echo canceling signal Ĉ·y(n−1) is a signal that depends on the canceling sound output from the speaker  22  and which reaches the microphone  18 . 
     The corrective value Ĉ represents signal transfer characteristics from an input terminal of the DAC  65  to an output terminal of the ADC  59 , including the transfer characteristics C from the speaker  22  to the microphone  18 . 
     The signal transfer characteristics actually are measured as follows: As shown in  FIG. 2 , a signal transfer characteristics measuring device  300 , which comprises a Fourier transforming device, is connected between the input and output terminals of the controller  202 . The signal transfer characteristics measuring device  300  measures signal transfer characteristics based on a test signal, which is input from the controller  202  to the DAC  65 , and a signal output from the subtractor  60  to the controller  202 . In  FIGS. 1 and 2 , the signal transfer characteristics measured by the signal transfer characteristics measuring device  300  are set as the corrective value Ĉ in the echo canceler  58 . Depending on how the signal transfer characteristics measuring device  300  measures the signal transfer characteristics, the corrective value Ĉ may represent signal transfer characteristics from the speaker  22  to the microphone  18 , or alternatively, signal transfer characteristics from the output terminal to the input terminal of the controller  202 , including signal transfer characteristics from the speaker  22  to the microphone  18 , which are measured as described above. 
     The corrective value (transfer characteristics) Ĉ including the transfer characteristics C is identified according to the aforementioned measuring process. As described above, the transfer characteristics C are divided into gain characteristics (amplitude change) G′ and a phase delay (phase characteristics) φ′. The corrective value Ĉ is divided into gain characteristics (amplitude change) G and a phase delay (phase characteristics) φ. 
     The subtractor  60  subtracts the echo canceling signal Ĉ·y(n−1), dependent on the canceling sound from the canceling error signal e(n), which in turn depends on the canceling error signal, thereby estimating a resonant noise d(n) at the position of the microphone  18 . Then, the subtractor  60  outputs a first basic signal x(n) based on the resonant noise d(n) to the controller  202 . Based on the input first basic signal x(n), the controller  202  generates a control signal y(n) depending on a canceling sound Ĉ·y(n), which is opposite in phase with and has the same amplitude as the resonant noise d(n+1) to be silenced, in a next sampling event (n+1) at the position of the microphone  18 . 
     The first fundamental concept will be described in more specific detail with reference to  FIG. 1  and  FIGS. 3A through 3C , showing vectors on a Gaussian plane. 
     In the sampling event n, the canceling sound that reaches the microphone  18  is expressed by Ĉ·y(n−1). Therefore, the canceling error signal e(n) output from the microphone  18  is expressed according to the following equation (3):
 
 e ( n )= d ( n )+Ĉ·y( n− 1)  (3)
 
     The ADC  59  converts the canceling error signal e(n) from an analog signal into a digital signal, and outputs the digital canceling error signal e(n) to the subtractor  60 . 
     The echo canceler  58  generates an echo canceling signal Ĉ·y(n−1) by correcting the control signal y(n−1) output from the controller  202  in the preceding sampling event (n−1) with the corrective value Ĉ, and outputs the echo canceling signal Ĉ·y(n−1) to the subtractor  60 . 
     The subtractor  60  subtracts the echo canceling signal Ĉ·y(n−1) from the canceling error signal e(n), thereby estimating the resonant noise d(n), and outputs a first basic signal x(n) based on the resonant noise d(n) to the controller  202 . 
     In view of the equation (3), the first basic signal x(n) is expressed according to the following equation (4):
 
 x ( n )= e ( n )− Ĉ·y ( n− 1)≈ d ( n )  (4)
 
     According to equation (4), the first basic signal x(n) corresponds to the resonant noise d(n) at the position of the microphone  18 , which is determined based on the canceling error signal e(n) and the control signal y(n). 
     Generation of the control signal y(n) in the controller  202  shall be described below. 
     As shown in  FIG. 3A , if the controller  303  can generate, in the sampling event n, a control signal y(n) (see  FIG. 3B ) depending on a canceling sound C·y(n) {=−d(n+1)}, which is opposite in phase with and has the same amplitude as the resonant noise d(n+1) to be silenced, in a next sampling event (n+1) at the position of the microphone  18 , based on the first basic signal x(n) {≈d(n)} in the present sampling event n, then when the speaker  22  outputs canceling sounds based on the control signal y(n) into the passenger compartment  14 , the resonant noise d(n+1) can reliably be silenced by the canceling sound Ĉ·y(n). 
     In other words, as long as resonant noise d(n) is present, the control signal y(n) can be output, so that the resonant noise d(n+1) at the position of the microphone  18  can stably be silenced. 
     As described above, the transfer characteristics C from the speaker  22  to the microphone  18  are identified by the corrective value Ĉ, and the corrective value Ĉ is divided into the gain characteristics G and the phase delay φ. Therefore, as shown in  FIG. 3B , the canceling sound Ĉ·y(n) that reaches the microphone  18  is generated by multiplying the magnitude of the control signal y(n) by G, and rotating G·y(n) through the phase delay φ. The controller  202  thus generates the control signal y(n) using the first basic signal x(n). 
     However, the control signal y(n) cannot be generated only from the first basic signal x(n) shown in  FIGS. 3A and 3B . 
     Consequently, as shown in  FIG. 3C , a second basic signal x′(n), which is orthogonal to and has the same amplitude as the first basic signal x(n), is generated and the control signal y(n) is generated based on the first basic signal x(n) and the second basic signal x′(n). In this case, the control signal y(n) is represented by a combined vector of A·x(n), which is the product of the first basic signal x(n) and the filter coefficient (gain) A, and B·x′(n), which is the product of the second basic signal x′(n) and the filter coefficient (gain) B {y(n)=A·x(n)+B·x′(n)}. 
     Specifically, as shown in  FIGS. 1 and 3C , the controller  202  regards the first basic signal x(n) as a cosine signal expressed according to the following equation (5):
 
 x ( n )=cos {2 πf×t ( n )}≈ d ( n )  (5)
 
     The delay filter  54  delays the first basic signal x(n) by a time Z −n  (90°) corresponding to a ¼ period of the resonant frequency f determined by the resonant characteristics of the passenger compartment  14 , thereby generating a cosine signal (second basic signal) x′(n) which is orthogonal to and has the same amplitude as the first basic signal x(n), as expressed according to the following equation (6): 
     
       
         
           
             
               
                 
                   
                     
                       
                         
                           
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     The first filter  62  generates a first corrective signal A·x(n) by multiplying the first basic signal x(n) by the filter coefficient A, and outputs the generated first corrective signal A·x(n) to the adder  56 . The adder  56  generates the control signal y(n) by combining the first corrective signal A·x(n) and the second corrective signal B·x′(n). The control signal y(n) is expressed according to the following equation (7): 
     
       
         
           
             
               
                 
                   
                     
                       
                         
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     When the DAC converts the control signal y(n) from a digital signal into an analog signal and the speaker  22  outputs a canceling sound based on the analog control signal y(n) into the passenger compartment  14 , the resonant noise d(n+1) at the position of the microphone  18  is reduced by the canceling sound Ĉ·y(n), which reaches the microphone  18  in the sampling event (n+1). As described above, the canceling sound Ĉ·y(n) is opposite in phase with the resonant noise d(n+1), and the product G·y(n) is a signal component produced by removing the phase delay φ from the canceling sound Ĉ·y(n). 
     According to the first fundamental concept, when the microphone  18  outputs the canceling error signal e(n), the control signal y(n) {=−d(n+1)/Ĉ}, which serves to cancel out the resonant noise d(n+1) to be silenced at the position of the microphone  18 , can be generated from the first basic signal x(n) and the second basic signal x′(n). Therefore, the canceling sound Ĉ·y(n) can be generated simply and accurately without the need for a FIR adaptive filter. Hence, the ANC  204  is simpler in arrangement and less expensive to manufacture. 
     Since the first basic signal x(n) is used as representing the resonant sound d(n) that is determined by subtracting the echo canceling signal Ĉ·y(n−1) from the canceling error signal e(n), the control signal y(n) can be generated as long as the resonant noise d(n) is present, so that the resonant noise d(n+1) at the position of the microphone  18  can be silenced stably. 
       FIGS. 4 and 5  illustrate a process of generating the second basic signal x′(n) with the delay filter  54 . It is assumed that the control unit  50  has a sampling period T. 
     The delay time Z −n , which corresponds to a ¼ period of the resonant frequency f as described above, is set to a value expressed as Z −n &gt;&gt;T and Z −n =m×T (m: an integer). For example, if the control frequency f=30 [Hz] and the sampling frequency (=1/T) is 3000 [Hz], then since Z −n =(¼)×( 1/30) [s]= 1/120[s] and T= 1/3000[s], m=Z −n /T=3000/120=25. One period ( 1/30[s]) for the control frequency f corresponds to 100 sampling events, with respect to T= 1/3000[s], and Z −n = 1/120[s] corresponds to 25 sampling events (a time depending on π/2). 
     In  FIG. 4 , the delay filter  54  (see  FIG. 1 ) comprises N (N=m+1) buffers. 
     In  FIG. 4 , it shall be assumed for the sake of brevity that m=4, N=m+1=5, i.e., the delay time Z −n  is four times the sampling period, and the delay filter  54  comprises five buffers. As described above, when the first basic signal x(n) is a cosine signal, the second basic signal x′(n) is a sine signal. Therefore, in  FIG. 4 , the first basic signal x(n) is represented by a cosine signal  220 , and the second basic signal x′(n) is represented by a sine signal  222 . 
     The delay filter  54  (see  FIG. 1 ) successively stores instantaneous values an (n=1, 2, . . . , i, . . . ), which are output as the cosine signal  220  from the subtractor  60  in respective sampling events, in the respective buffers  0  through  4 . 
     Since the delay time Z −n =m·T=4T, the delay filter  54  reads a stored instantaneous value a(i−4) from a buffer, which stores the instantaneous value a(i−4) that is m sampling events (n=i−m) prior to the buffer storing an instantaneous value ai, and outputs the read instantaneous value a(i−4) as a second basic signal x′(n) in the sampling event i. For example, in the sampling event i=7, an instantaneous value a 7  is stored in the buffer  1 , and an instantaneous value a 3 , which is stored in the buffer  2  and which is four sampling events (n=3) prior to the buffer  1 , is read and output as a second basic signal x′( 7 ) in the sampling event i=7. 
     Therefore, if the first basic signal x(i) is represented by an instantaneous value ai output from the subtractor  60  at the timing of the sampling event i, then the second basic signal x′(n) is represented by an instantaneous value of a(i−4), which is delayed by a ¼ period from the first basic signal x(i). 
     The number of buffers is given as N=m+1 for storing instantaneous value data an corresponding to the delay time Z −n , and also for storing the instantaneous value an, which is output from the subtractor  60  during the present sampling event n. 
     As shown in  FIG. 4 , since the number of buffers is one greater than m, the buffer storing the instantaneous value a(i−4), which is m sampling events (n=i−m) prior to the buffer storing the instantaneous value ai in the sampling event n=i, refers to a buffer that is updated during a next sampling event (i+1). 
     If the delay filter  54  comprises a shift register instead of buffers, then the shift register comprises N=m registers. 
     In this case, in the respective sampling events n, the delay filter  54  successively stores instantaneous values an in the respective registers, and reads the oldest instantaneous value (oldest data) an prior to being stored as a second basic signal x′(n). If the first basic signal x(i) is represented by the instantaneous value an output from the subtractor  60  at the timing of the sampling event i, then the second basic signal x′(n) is represented by a(i−4) and is delayed by a ¼ period from the first basic signal x(i). 
     According to the first fundamental concept, as described above, when the microphone  18  outputs the canceling error signal e(n), the control signal y(n) {=−d(n+1)/Ĉ}, which acts to cancel out the resonant noise d(n+1) to be silenced at the position of the microphone  18 , can be generated from the first basic signal x(n) and the second basic signal x′(n). Therefore, the canceling sound Ĉ·y(n) can simply and accurately be generated, without the need for a FIR adaptive filter. Hence, the ANC  204  is simpler in arrangement and less expensive to manufacture. 
     Since the first basic signal x(n) is used to represent the resonant sound d(n) that is determined by subtracting the echo canceling signal Ĉ·y(n−1) from the canceling error signal e(n), the control signal y(n) can be generated as long as the resonant noise d(n) is present, so that the resonant noise d(n+1) at the position of the microphone  18  can be silenced stably. 
     The second fundamental concept will be described below with reference to  FIGS. 6 and 7 . The second fundamental concept differs from the first fundamental concept (see  FIGS. 1 through 5 ), in that a controller  202  comprises a delay filter  55 , an adder  56 , and a filter (amplitude adjuster)  70  having a predetermined filter coefficient (gain) P. 
     The second fundamental concept is similar to the first fundamental concept, in that the control signal y(n) depending on the canceling sound Ĉ·y(n), which is in opposite phase with and has the same amplitude as the resonant noise d(n+1) to be silenced in the next sampling event (n+1) at the position of the microphone  18 , is generated in the present sampling event n based on the first basic signal x(n) {≈d(n)} in the sampling event n. However, the second fundamental concept differs from the first fundamental concept as to how the control signal y(n) is generated in the controller  202 . According to the second fundamental concept, the corrective value Ĉ is also divided into gain characteristics (amplitude change) G and a phase delay (phase characteristics) φ. 
     The delay filter  55  generates a second basic signal x″(n) expressed according to the following equation (8) by delaying the first basic signal x(n) expressed according to the above equation (5) by a predetermined time Z −m  (thereby delaying the phase thereof by a predetermined angle 2Ψ):
 
 x ″( n )=cos [2 πf×{t ( n )+2Ψ}]  (8)
 
     Therefore, as shown in  FIG. 7 , the second basic signal x″(n) is a signal that has the same amplitude as the first basic signal x(n) while being 2Ψ out of phase with the first basic signal x(n). 
     The predetermined time Z −m  has a value based on the control frequency f, which is equal to the resonant frequency f of the resonant noise d(n), and a phase delay (phase characteristics) φ of the transfer characteristics (corrective value) Ĉ of the sound at the control frequency f. Specifically, the predetermined time Z −m  is a time corresponding to the phase value 2Ψ, which is twice the value that is produced by subtracting the phase delay (phase characteristics) φ from the phase difference between the first basic signal x(n) and the canceling sound Ĉ·y(n), which is opposite in phase with and has the same amplitude as the resonant noise d(n+1). The predetermined time Z −m  actually is determined on a trial and error basis, based on the gain P of the filter  70  and a phase value Ψ at the time a test noise having the control frequency f is generated in the passenger compartment  14 , wherein the generated test noise is silenced at the position of the microphone  18 . 
     The adder  56  adds the first basic signal x(n) and the basic signal x″(n) into a combined signal {x(n)+x″(n)}. The adder  56  outputs the combined signal {x(n)+x″(n)} to the filter  70 . 
     Based on the combined signal {x(n)+x″(n)} from the adder  56 , the filter  70  generates a control signal y(n). 
     Specifically, as shown in  FIG. 7 , the filter  70  multiplies the first basic signal x(n) by the filter coefficient (gain) P in order to generate a product signal P·x(n), multiplies the second basic signal x″(n) by the filter coefficient (gain) P so as to generate a product signal P·x″(n), and combines the product signal P·x(n) and the product signal P·x″(n) into the control signal y(n). 
     The control signal y(n) and the first basic signal x(n) make up a triangle  206 , whereas the control signal y(n) and the second basic signal x″(n) make up a triangle  208 . Since the triangles  206 ,  208  have equal sides along the control signal y(n), equal sides (P) along the basic signals x(n), x′(n)k, and equal phase values Ψ, the triangles  206 ,  208  are congruent, because the pairs of corresponding sides and the included angle thereof are both equal. Accordingly, the control signal y(n) is expressed according to the following equation (9): 
     
       
         
           
             
               
                 
                   
                     
                       
                         
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     Therefore, the filter  70  generates the control signal y(Nn) by multiplying {x(n)+x″(n)} by the filter coefficient (gain) P. 
     According to the second fundamental concept, as described above, when the microphone  18  outputs the canceling error signal e(n), the control signal y(n) (=−d(n+1)/Ĉ), which acts to cancel out the resonant noise d(n+1) to be silenced at the position of the microphone  18 , can be generated from the first basic signal x(n) and the second basic signal x″(n). Therefore, the canceling sound Ĉ·y(n) can simply and accurately be generated without the need for a FIR adaptive filter. Hence, the ANC  204  is simpler in arrangement and less expensive to manufacture. 
     Specific examples of the ANC  204  (ANCs  10 A through  10 H according to first through eighth embodiments of the present invention) based on the first and second fundamental concepts (see  FIGS. 1 through 7 ) shall be described below with reference to  FIGS. 8 through 17 . In each of these embodiments, parts which are identical to those of the first and second fundamental concepts are denoted using identical reference characters, and such parts will not be described in detail below. 
       FIGS. 8 and 9  show in block form an ANC  10 A according to a first embodiment of the present invention, which is a specific example of the first fundamental concept (see FIGS.  1  through  5 ). 
     The ANC  10 A is incorporated in a vehicle  12  as shown in  FIG. 8 . The ANC  10 A basically comprises an ANC electronic controller  20  including a microcomputer  52  (see  FIG. 9 ), a speaker  22  disposed in a given position in the vehicle  12 , e.g., below a front seat  24 , and a microphone  18  disposed near the position of an ear of a passenger, not shown, in a passenger compartment  14  of the vehicle  12 , e.g., near the headrest  26  of the front seat  24 . 
     The noise at the position of the microphone  18  includes (1) a noise generated in the passenger compartment  14  by vibrations of an engine (not shown) or the like in the vehicle  12 , and a noise generated by a noise source, and a periodic noise {engine muffled sound (engine noise)} generated in the passenger compartment  14  by the above vibrations, and by vibrations of the noise source, and (2) an aperiodic low-frequency noise (drumming noise (road noise)) generated in the passenger compartment  14  due to contact between plural tires  19  and the road  21  while the vehicle  12  is running. 
     The road noise (2) is produced as a resonant sound (the resonant noise d(n) described above) having a high sound pressure level at a certain resonant frequency f due to the resonant characteristics in the passenger compartment  14 . The resonant sound is a road noise having a central frequency equal to the resonant frequency f of 40 [Hz], for example. Specifically, the resonant sound refers to road noises that resonate within the passenger compartment  14  at the resonant frequency f, which is determined by the structure of the resonant chamber, i.e., the transverse and longitudinal dimensions of the passenger compartment  14 . If the vehicle  12  is a passenger automobile, such as a sedan or the like, then the passenger compartment  14  has resonant characteristics represented by an acoustic mode in which the resonant sounds resonate at a frequency of about 40 [Hz] in the passenger compartment  14 . Therefore, the resonant frequency f is a known frequency, which can be determined by the structure of the passenger compartment  14 . 
     Since the road noise is strongly affected by the acoustic mode of the passenger compartment  14 , the microphone  18  may be located in the passenger compartment  14  at an antinode  16   a  (an area in front of the front seat  24  in the passenger compartment  14 ) of the acoustic mode thereof. The acoustic mode also has other antinodes, including an antinode  16   b  extending between the front seat  24  and a rear seat  36 , and an antinode  16   c  extending above the rear seat  36  and a trunk compartment  38  behind the rear seat  36 . In order to detect road noises at the antinodes  16   a  through  16   c , (1) other microphones  30 ,  32 ,  34  may be disposed near the roof  28 , i.e., in a roof lining, not shown, provided on the inner surface of the roof  28 , (2) a microphone  40  may be disposed near a lower end of the front seat  24  at the feet of the passenger seated in the front seat  24 , and (3) a microphone  42  may be disposed in the trunk compartment  38 . Accordingly, the microphones  30 ,  32 ,  34 ,  40 , and  42  can output canceling error signals e(n) to the ANC electronic controller  20 . 
     In addition, another speaker  44  may be disposed in a rear tray  43  behind the rear seat  36 , for outputting a canceling sound. 
     In the following description, it will be assumed that only the microphone  18  and the speaker  22  are disposed in the passenger compartment  14 . 
     As shown in  FIG. 2 , the ANC electronic controller  20  includes a control unit  50 , a low-pass filter (LPF)  66  for passing and outputting a signal having a predetermined frequency or lower, from the canceling error signal e(n) output from the microphone  18 , and an LPF  68  for passing and outputting, to the speaker  22 , a signal having a predetermined frequency or lower, from the control signal y(n) output from the control unit  50 . The control unit  50  has a sampling period set to a given period (e.g., 1/3000 [s]), which is much shorter than the delay time, e.g., 1/160 [s], of the delay filter  54 . 
     The echo canceler  58  comprises a FIR filter or a notch filter having a fixed filter coefficient. 
     The LPF  66  comprises an antialiasing filter for removing folding noises having a predetermined frequency {a frequency higher than the control frequency f of the control signal y(n)} or higher from the canceling error signal e(n) input from the microphone  18 . The LPF  66  then supplies the canceling error signal e(n) to the microcomputer  52 . 
     The LPF  68  comprises a reconstruction filter for removing from the control signal y(n) signal components having frequencies higher than the control frequency f and which are generated when the control signal y(n) is converted into an analog signal by the DAC  65 . The LPF  68  then outputs the control signal y(n), from which the high-frequency components have been removed, to the speaker  22 . 
     Since the control unit  50  of the ANC  10 A is capable of generating the control signal y(n) through a simpler digital signal processing method, the computational burden for generating the control signal y(n) is reduced. Further, since the control unit  50  consists of a simple arrangement using the microcomputer  52 , which is relatively inexpensive, the ANC  10 A can be manufactured inexpensively. As a result, the ANC  10 A may be reduced in overall unit size, and can be combined with a digital audio unit in the vehicle  12 . 
     Furthermore, since the LPF  66  comprises an antialiasing filter, although the control unit  50  is functionally realized by the microcomputer  52 , which generates the control signal y(n) according to a digital signal processing method, the LPF  66  removes folding noises having a predetermined frequency or higher from the canceling error signal e(n), and then supplies the canceling error signal e(n) to the microcomputer  52 . Accordingly, the control signal y(n) can be generated accurately in the microcomputer  52 . 
     In addition, since the LPF  68  comprises a reconstruction filter, although the control unit  50  is functionally realized by the microcomputer  52 , which generates the control signal y(n) according to a digital signal processing method, converts the control signal y(n) into an analog signal, and outputs the analog control signal Y(n) to the speaker  22 , the LPF  68  removes high-frequency components from the analog control signal y(n), so that the analog control signal y(n) possesses a smooth waveform over time. As a result, the speaker  22  can output a high-quality canceling sound based on the control signal y(n), from which high-frequency components have been removed. 
     An ANC  10 B according to a second embodiment, which is a specific example of the second fundamental concept (see  FIGS. 6 and 7 ), will be described below with reference to  FIG. 10 . 
     The ANC  10 B includes the filter  70  described above with reference to  FIGS. 6 and 7 . Therefore, the ANC  10 B has one filter fewer than the filters used in the ANC  10 A. As a result, the computational burden on the ANC  10 B in generating the control signal y(n) is further reduced. 
     An ANC  10 C according to a third embodiment will be described below with reference to  FIGS. 11 and 12 . 
     The ANC  10 C differs from the ANC  10 A (see  FIG. 9 ) according to the first embodiment, in that a bandpass filter (BPF)  72  is connected to the input side of the microcomputer  52 . 
     From the canceling error signal e(n) output from the LPF  66 , the BPF  72  passes and outputs, to the microcomputer  52 , only a signal within a predetermined frequency band, having a central frequency equal to the control frequency of 40 [Hz], for example, of the control signal y(n). In other words, from the canceling error signal e(n), the BPF  72  passes only a signal corresponding to a road noise (resonant sound) having a central frequency of about 40 [Hz], and outputs the signal through the ADC  59  to the microcomputer  52 . 
       FIG. 12  shows sound pressure vs. frequency characteristics of a noise at the position of the microphone  18  (see  FIG. 12 ).  FIG. 12  illustrates a comparison between a characteristic curve plotted when a silencing control mode is carried out (CONTROLLED) at the position of the microphone  18  by the ANC  10 C, for outputting the canceling sound from the speaker  22  into the passenger compartment  14 , and a characteristic curve plotted when a silencing control mode is not carried out (NOT CONTROLLED) at the position of the microphone  18  by the ANC  10 C. In the silencing control mode that is carried out (CONTROLLED), the control frequency f of the control signal y(n) is set at 40 [Hz]. 
     It can be seen from  FIG. 11  that when the silencing control mode is carried out (CONTROLLED), the noise (road noise) at the position of the microphone  18  is reliably lowered within the frequency band from 30 [Hz] to 50 [Hz] around 40 [Hz]. 
     The ANC  10 C according to the third embodiment offers the same advantages as those of the ANC  10 A (see  FIG. 9 ) according to the first embodiment described above. In addition, although the control unit  50  is functionally realized by the microcomputer  52  for generating the control signal y(n) according to a digital signal processing method, since, from the canceling error signal e(n), the BPF  72  passes only a signal inside of a predetermined frequency band (a frequency band having a central frequency of 40 [Hz]), and then supplies the signal to the microcomputer  52 , the microcomputer  52  can generate the control signal y(n) more accurately. 
     An ANC  10 D according to a fourth embodiment will be described below with reference to  FIG. 13 . 
     The ANC  10 D differs from the ANC  10 C (see  FIG. 11 ) according to the third embodiment, in that the ANC electronic controller  20  includes an allpass filter (APF)  74 , instead of the delay filter  54 , disposed outside of the microcomputer  52 . The ANC  10 D also includes a DAC (delay filter DAC)  75  and an ADC (delay filter ADC)  77 . 
     The DAC  75  converts the first basic signal x(n) from a digital signal into an analog signal, and outputs the analog first basic signal x(n) to the APF  74 . 
     The APF  74  comprises a delay filter having a phase delay at the control frequency f of the control signal y(n), which is set to a phase delay (90°) corresponding to a ¼ period of the control frequency f. Therefore, the APF  74  shifts the first basic signal x(n) input from the DAC  75  in phase by 90°, thereby generating a second basic signal x′(n), and outputs the second basic signal x′(n) to the ADC  77 . 
     The ADC  77  converts the second basic signal x′(n) from an analog signal into a digital signal, and outputs the digital second basic signal x′(n) to the second filter  64 . 
     The ANC  10 D according to the fourth embodiment offers the same advantages as those of the ANC  10 C (see  FIG. 11 ) according to the third embodiment described above. In addition, since the delay filter comprises the APF  74 , which is in the form of an analog circuit, the APF  74  does not need to be included in the microcomputer  52 . Hence, the microcomputer  52  may be of a simpler design. 
     An ANC  10 E according to a fifth embodiment will be described below with reference to  FIG. 14 . 
     In  FIG. 9 , an echo canceling signal Ĉ·y(n) is generated by multiplying, by the corrective value Ĉ, the control signal y(n), which is generated by combining the first corrective signal A·x(n) that is produced by multiplying the first basic signal x(n) by the filter coefficient (gain) A, and the second corrective signal B·x′(n) that is produced by multiplying the second basic signal x′(n) by the filter coefficient (gain) B [Ĉ·y(n)=Ĉ{A·x(n)+B·x′(n)}]. 
     The echo canceling signal Ĉ·y(n) also can be generated by multiplying the first basic signal x(n) by the corrective value Ĉ, and thereafter by multiplying the product by the filter coefficient A, multiplying the second basic signal x′(n) by the corrective value Ĉ, and thereafter by multiplying the product by the filter coefficient B, and finally combining A·Ĉ·x(n) and B·Ĉ·x′(n) [A·Ĉ·x(n)+B·Ĉ·x′(n)=Ĉ{A·x(n)+B·x′(n)}=Ĉ·y(n)]. 
     Based on the latter alternative, it is possible to generate the echo canceling signal Ĉ·y(n) according to a method of generating the first basic signal and the second basic signal at the position of the microphone  18 , as disclosed in Japanese Laid-Open Patent Publication No. 2004-361721. 
     Specifically, the product of the first basic signal x(n) as a cosine signal and the corrective value Ĉ represents a first basic signal at the position of the microphone  18 . The product of the second basic signal x′(n) as a sine signal and the corrective value Ĉ represents a second basic signal at the position of the microphone  18 . 
     If a cosine corrective value based on the cosine value of the phase delay φ of the corrective value Ĉ is represented by Cr, and a sine corrective value based on the sine value of the phase delay φ of the corrective value Ĉ is represented by Ci, then the first basic signal at the position of the microphone  18  is expressed as a signal generated by subtracting the product Ci·x′(n) of the sine corrective value Ci and the second basic signal x′(n) from the product Cr·x(n) of the cosine corrective value Cr and the first basic signal x(n), i.e., a differential signal Sm {Sm=Cr·x(n)−Ci·x′(n)}. The second basic signal at the position of the microphone  18  is expressed as a signal generated by adding the product Cr·x′(n) of the cosine corrective value Cr and the second basic signal x′(n) to the product Ci·x(n) of the sine corrective value Ci and the first basic signal x(n), i.e., a sum signal Sp {Sp=Cr·x′(n)+Ci·x(n)}. 
     Therefore, the echo canceling signal Ĉ·y(n) is generated by adding the product A·Sm of the differential signal Sm and the filter coefficient A to the product B·Sp of the sum signal Sp and the filter coefficient B. 
     More specifically, as shown in  FIG. 14 , the echo canceler  58  comprises a first cosine corrector  80  and a second cosine corrector  84  each having the cosine corrective value Cr, a first sine corrector  82  and a second sine corrector  86  each having the sine corrective value Ci, a subtractor  88 , a first adder  90 , a first correcting filter  92  having the same filter coefficient (gain) A as the first filter  62 , a second correcting filter  94  having the same filter coefficient (gain) B as the second filter  64 , and a second adder  96 . 
     The first cosine corrector  80  corrects the first basic signal x(n) with the cosine corrective value Cr, and then outputs the corrected signal Cr·x(n) to the subtractor  88 . The first sine corrector  82  corrects the second basic signal x′(n) with the cosine corrective value Cr, and then outputs the corrected signal Cr·x′(n) to the subtractor  88 . The second cosine corrector  84  corrects the second basic signal x′(n) with the cosine corrective value Cr, and then outputs the corrected signal Cr·x′(n) to the first adder  90 . The second sine corrector  86  corrects the first basic signal x(n) with the sine corrective value Ci, and then outputs the corrected signal Ci·x(n) to the first adder  90 . 
     The subtractor  88  subtracts the corrected signal Cr·x′(n) output from the first sine corrector  82  from the corrected signal Cr·x(n) output from the first cosine corrector  80 , thereby generating the differential signal Sm. The first adder  90  adds the corrected signal Cr·x′(n) output from the second cosine corrector  84  to the corrected signal Ci·x(n) output from the second sine corrector  86 , thereby generating the sum signal Sp. 
     The first correcting filter  92  corrects the differential signal Sm with the gain A, and outputs the corrected signal A·Sm to the second adder  96 . The second correcting filter  94  corrects the sum signal Sp with the gain B, and outputs the corrected signal B·Sp to the second adder  96 . 
     The second adder  96  adds the corrected signal A·Sm output from the first correcting filter  92  to the corrected signal B·Sp output from the second correcting filter  94 , thereby generating an echo canceling signal Ĉ·y(n), and outputs the echo canceling signal Ĉ·y(n) in accordance with the timing of a sampling event (n+1). 
     The ANC  10 E according to the fifth embodiment offers the same advantages as those of the ANC  10 C (see  FIG. 11 ) according to the third embodiment described above. In addition, the processing sequence for generating the echo canceling signal comprises a total of nine processes including arithmetic operations, i.e., four correcting processes carried out respectively by the first cosine corrector  80 , the second cosine corrector  84 , the first sine corrector  82 , and the second sine corrector  86 , one subtracting process carried out by the subtractor  88 , one adding process carried out by the first adder  90 , two correcting processes carried out respectively by the first correcting filter  92  and the second correcting filter  94 , and one adding process carried out by the second adder  96 . As a result, the amount of processing operations for generating the echo canceling signal is reduced. In other words, the echo canceling signals Ĉ·y(n−1), Ĉ·y(n) can be generated by a simpler arrangement, without the need for a FIR filter. 
     An ANC  10 F according to a sixth embodiment will be described below with reference to  FIG. 15 . 
     The ANC  10 F according to the sixth embodiment differs from the ANC  10 E (see  FIG. 14 ) according to the fifth embodiment, in that the ANC electronic controller  20  includes the APF  74 , which is used as a delay filter. 
     The ANC  10 F offers the same advantages provided by the APF  74  of the ANC  10 D (see  FIG. 13 ) according to the fourth embodiment, as well as the advantages of the ANC  10 E (see  FIG. 14 ) according to the fifth embodiment. 
     An ANC  10 G according to a seventh embodiment will be described below with reference to  FIG. 16 . 
     The ANC  10 G differs from the ANC  10 E (see  FIG. 14 ) according to the fifth embodiment, in that the microcomputer  52  (the control unit  50 ) includes a first filter coefficient updater  100  and a second filter coefficient updater  102 , each of which comprises a least mean square algorithm (LMS) operator. Further, each of the first filter  62 , the second filter  64 , the first correcting filter  92 , and the second correcting filter  94  comprises an adaptive filter, or more preferably an adaptive notch filter. 
     The first filter coefficient updater  100  performs an adaptive processing sequence for updating the filter coefficients A of the first filter  62  and the first correcting filter  92  in order to minimize the canceling error signal e(n) based on the differential signal Sm and the canceling error signal e(n), i.e., a processing sequence for calculating the filter coefficients A so as to minimize the canceling error signal e(n) based on the least mean square algorithm. 
     The second filter coefficient updater  102  performs an adaptive processing sequence for updating the filter coefficients B of the second filter  64  and the second correcting filter  94 , so as to minimize the canceling error signal e(n) based on the sum signal Sp and the canceling error signal e(n). 
     The ANC  10 G according to the seventh embodiment offers the same advantages as those of the ANC  10 E (see  FIG. 14 ) according to the fifth embodiment described above. In addition, even if the transfer characteristics C and the corrective value Ĉ vary due to mass-production-induced variations in the layout of the speaker  22  and the microphone  18  in the passenger compartment  14 , or undergo changes due to aging or the like, since the filter coefficients A of the first filter  62  and the first correcting filter  92  as well as the filter coefficients B of the second filter  64  and the second correcting filter  94  are updated under an adaptive control, noise inside the passenger compartment  14  can still be silenced accurately. 
     An ANC  10 H according to an eighth embodiment will be described below with reference to  FIG. 17 . 
     The ANC  10 H differs from the ANC  10 G (see  FIG. 16 ) according to the seventh embodiment, in that the ANC electronic controller  20  includes the APF  74  for use as a delay filter. 
     The ANC  10 H offers the advantages provided by both the APF  74  of the ANC  10 D (see  FIG. 13 ) according to the fourth embodiment, as well as the advantages of the ANC  10 G (see  FIG. 16 ) according to the seventh embodiment. 
     Although certain preferred embodiments of the present invention have been shown and described in detail, it should be understood that various changes and modifications may be made therein without departing from the scope of the invention as set forth in the appended claims.