Patent Publication Number: US-9433055-B2

Title: Lighting device and illumination apparatus including same

Description:
FIELD OF THE INVENTION 
     The present invention relates to a lighting device and an illumination apparatus including same. 
     BACKGROUND OF THE INVENTION 
     Recently, there is an increasing consumer interest in illumination and an illumination apparatus using light emitting diodes (LED elements) as light sources are being diversified. Under these circumstances, there is an increasing number of high-power products and the like in which LED modules, each having multiple LED elements connected in series to each other, are connected in parallel. Further, in order to cope with the wide variability of LEDs, a constant current circuit for supplying a constant current may be provided in the LED modules connected in parallel. 
     However, in a case where the LED modules are connected in parallel, if some of the LED modules are detached or an open-circuit mode failure occurs therein, there may flow concentrated currents through the other LED modules, and it may lead to destruction and degradation of the LED modules. Even in a case where the current supplied to the entire load is controlled to be constant by using the constant current circuit, the concentrated currents may flow through some of the LED modules. Accordingly, it has been necessary to establish a measure for each LED module. 
     Thus, there is an illumination apparatus in which a constant current circuit and a connection state detection circuit are provided for each of LED modules connected in parallel (see, e.g., Japanese Patent Application Publication No. 2009-21175). In this illumination apparatus, if it is detected that a certain LED module is detached, the supply of the current to the corresponding LED module is stopped, thereby preventing concentrated currents from flowing through the other LED modules. 
     Further, there is a lighting circuit which detects an abnormality in an LED load and safely turns on a light source of a vehicle lamp (see, e.g., Japanese Patent Application Publication No. 2004-134147). The lighting circuit supplies a constant current to the entire light source having LED loads connected in parallel. Further, a sense resistor is connected in series to each LED load, and an abnormality such as failure or detachment of an LED load is detected by sensing a voltage across each sense resistor. Further, if an abnormality is detected, the power supplied to the entire LED load is reduced by adjusting a drive signal of a switching regulator, thereby maintaining a safe operation. 
     However, in the illumination apparatus of Japanese Patent Application Publication No. 2009-21175, since the constant current circuits and the connection state detection circuits need to be provided in the same number as the number of the LED modules connected in parallel, the circuit configuration becomes complicated, and it results in a large power loss due to the constant current circuits and the connection state detection circuits and a low conversion efficiency of the illumination apparatus. 
     Further, in the illumination apparatus of Japanese Patent Application Publication No. 2004-134147, since the sense resistors need to be provided in the same number as the number of the LED loads connected in parallel, it results in a large power loss due to the sense resistors and a low conversion efficiency of the lighting circuit. 
     SUMMARY OF THE INVENTION 
     In view of the above, the present invention provides a lighting device capable of reducing a power loss and preventing concentrated currents from flowing through normally operating light emitting modules when an abnormality develops in the load, and an illumination apparatus including same. 
     In accordance with an embodiment of the present invention, there is provided a lighting device including: a lighting unit which controls a current being supplied to a load, in which light emitting modules, each having one or more semiconductor light emitting elements connected in series, are connected in parallel, to be a constant current; a current detector which detects a current flowing through one of the light emitting modules; and an abnormality detector which compares a detected value from the current detector with an upper limit and a lower limit of a predetermined current range to detect an abnormality in the load. The abnormality detector detects the abnormality in the load if the detected value from the current detector is larger than the upper limit or smaller than the lower limit, and if the abnormality detector detects the abnormality in the load, the lighting unit reduces the current being supplied to the load. 
     Further, if the abnormality detector detects the abnormality in the load, the lighting unit may perform an intermittent operation for intermittently reducing the current being supplied to the load, and if the abnormality detector is switched from a state in which the abnormality in the load is detected to a state in which the abnormality in the load is not detected while the lighting unit performs the intermittent operation, the lighting unit may stop the intermittent operation. 
     Further, as a difference between the upper limit of the predetermined current range and the detected value from the current detector that is larger than the upper limit, or a difference between the lower limit of the predetermined current range and the detected value from the current detector that is smaller than the lower limit increases, the lighting unit may increase a reduction in the current being supplied to the load. 
     Further, the lighting unit may include a direct current (DC) power supply for outputting a DC power and a constant current supply unit for controlling the current being supplied to the load to be a constant current by using the DC power supply as an input power supply. 
     Further, the current detector may detect a current flowing through only said one of the light emitting modules. 
     In accordance with another embodiment of the present invention, there is provided an illumination apparatus including: the lighting device described in claim  1  or  2 ; and a load, in which light emitting modules, each having one or more semiconductor light emitting elements connected in series, are connected in parallel, and to which a current is supplied from the lighting device. 
     In accordance with the present invention, it is possible to reduce power loss by a simple configuration, and to prevent a concentrated current from flowing through normally operating light emitting modules when there develops an abnormality in the load. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The objects and features of the present invention will become apparent from the following description of embodiments, given in conjunction with the accompanying drawings, in which: 
         FIG. 1  illustrates a block diagram showing a configuration of a lighting device in accordance with a first embodiment of the present invention; 
         FIG. 2  illustrates a circuit diagram showing the configuration of the lighting device in accordance with the first embodiment of the present invention; 
         FIG. 3  illustrates a circuit diagram showing a configuration of an abnormality detector of the lighting device in accordance with the first embodiment of the present invention; 
         FIG. 4  illustrates a circuit diagram showing another configuration of the abnormality detector of the lighting device in accordance with the first embodiment of the present invention; 
         FIGS. 5A to 5E  illustrate circuit diagrams showing configuration examples of a step-down converter of the lighting device in accordance with the first embodiment of the present invention; 
         FIG. 6  illustrates a block diagram showing a configuration of a lighting device in accordance with a second embodiment of the present invention; and 
         FIG. 7  schematically shows an illumination apparatus in accordance with a third embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings, which form a part hereof. 
     First Embodiment 
       FIG. 1  illustrates a block diagram showing a configuration of a lighting device  1  in accordance with a first embodiment of the present invention. The lighting device  1  of this embodiment includes a filter circuit  2 , a rectifier circuit  3 , a step-up chopper circuit  4 , a step-down converter  5 , a control power supply circuit  6 , a current detector  7 , a step-up chopper controller  8 , a step-down converter controller  9 , a dimming controller  10 , and an abnormality detector  11 . 
     Each part of the lighting device  1  of this embodiment will be described with reference to a circuit diagram shown in  FIG. 2 . 
     A commercial AC power source  200  (e.g., 100 V, 50/60 Hz) is connected between input terminals of the filter circuit  2  via a connector CN 1 . A fuse F 1  is provided between the connector CN 1  and the filter circuit  2 . A parallel circuit of a varistor (surge voltage protection element) ZNR 1  and a filter capacitor C 1  is connected between the input terminals of the filter circuit  2 . A common mode choke coil (line filter) Lf 1  is connected to each input terminal of the filter circuit  2 . As the filter circuit  2  is configured as described above, it is possible to reduce a noise component in the input terminal. 
     The rectifier circuit  3  includes a full-wave rectifier DB 1  to which the output of the filter circuit  2  is inputted to full-wave rectify an AC voltage applied from the commercial AC power source  200  and a capacitor C 2  for high frequency bypass. As the rectifier circuit  3  is configured as described above, it is possible to full-wave rectify the AC power supplied from the commercial AC power source  200  and generate a ripple voltage at both terminals of the capacitor C 2 . 
     Further, a negative electrode of the DC output terminal of the full-wave rectifier DB 1  serves as a ground on a circuit board, and is high frequency grounded to a chassis potential FG through a series circuit of capacitors C 3  and C 4 . Hereinafter, a portion having the same potential as the negative electrode of the full-wave rectifier DB 1  is referred to as a circuit ground. 
     Main components of the step-up chopper circuit  4  include an inductor L 1 , a switching element Q 1 , a diode D 1  and a smoothing capacitor C 5 . Although the step-up chopper controller  8  is included in the step-up chopper circuit  4  in  FIG. 2  for convenience of illustration, the step-up chopper controller  8  is not a component of the step-up chopper circuit  4 . 
     Specifically, a series circuit including the inductor L 1 , the diode D 1  and the smoothing capacitor C 5  is connected between the DC output terminals of the full-wave rectifier DB 1 . A positive electrode of the DC output terminal of the full-wave rectifier DB 1  is connected to an anode of the diode D 1  through the inductor L 1 , and a cathode of the diode D 1  is connected to a positive electrode of the smoothing capacitor C 5 . Further, a series circuit including the switching element Q 1  containing an n channel MOSFET and a current detection resistor R 1  is connected between the circuit ground and a connection node between the inductor L 1  and the diode D 1 . 
     The switching element Q 1  has a drain connected to the anode of the diode D 1 , a source connected to the circuit ground through the resistor R 1 , and a gate connected to the step-up chopper controller  8  that will be described later. 
     In the step-up chopper circuit  4  configured as described above, the switching element Q 1  is controlled to be switched at a high frequency by the step-up chopper controller  8 . Accordingly, the step-up chopper circuit  4  steps up the ripple voltage outputted from the rectifier circuit  3  to generate a DC voltage (e.g., 410 V) smoothed by the smoothing capacitor C 5 . 
     The smoothing capacitor C 5  is a large-capacity capacitor including an aluminum electrolytic capacitor or the like, and a small-capacity capacitor C 6  for high frequency bypass is connected in parallel to the smoothing capacitor C 5 . The capacitor C 6  includes a film capacitor to bypass a high frequency component flowing through the smoothing capacitor C 5 . 
     Next, the step-up chopper controller  8  will be described. The step-up chopper controller  8  includes a power factor correction (PFC) circuit IC 1  and its peripheral circuits, and performs switching control of the switching element Q 1 . Further, the filter circuit  2 , the rectifier circuit  3 , the step-up chopper circuit  4  and the step-up chopper controller  8  correspond to a DC power supply described in the claims. 
     The PFC circuit IC 1  of this embodiment uses an IC chip of L6562A manufactured by STMicroelectronics (STME), which includes a first pin P 11  to an eighth pin  818 . Hereinafter, the function and operation of the first pin  811  to the eighth pin  818  will be described. 
     The eighth pin  818  (Vcc) is a power supply terminal and the sixth pin P 16  (GND) is a ground terminal. A control power supply voltage Vcc (hereinafter, referred to as a control voltage Vcc) outputted from the control power supply circuit  6  that will be described later is supplied between the eighth pin P 18  and the sixth pin P 16 . The PFC circuit IC 1  is driven by using the control voltage Vcc as an input power supply. Further, a capacitor C 11  is connected between the eighth pin P 18  and the sixth pin P 16 . The capacitor C 11  is a small-capacity capacitor for power supply bypass to remove noise from the control voltage Vcc. 
     The seventh pin P 17  (GD) is a gate drive terminal, and a series circuit including resistors R 14  and R 15  is connected between the seventh pin P 17  and the circuit ground. Further, a connection node between the resistor R 14  and the resistor R 15  is connected to a gate of the switching element Q 1 . Further, a series circuit including a resistor R 16  and a diode D 2  is connected in parallel to the resistor R 14 . An anode of the diode D 2  is connected to the gate of the switching element Q 1 . 
     Further, if the output level of the seventh pin P 17  becomes a high level, the current flows into the resistor R 15  through the resistor R 14  so that the voltage across the resistor R 15  increases. Further, if the voltage across the resistor R 15  becomes equal to or larger than a gate-source threshold voltage of the switching element Q 1 , the switching element Q 1  is turned on. Further, if the output level of the seventh pin P 17  becomes a low level, the charges accumulated between the gate and source of the switching element Q 1  are discharged through the diode D 2  and the resistor R 16 , so that the switching element Q 1  is turned off. 
     The fourth pin P 14  (CS) is a chopper current detection terminal to detect the current flowing through the switching element Q 1  by detecting the voltage across the current detection resistor R 1  through a noise filter circuit including a resistor R 12  and a capacitor C 10 . Further, if the detection value is equal to or larger than a threshold value, the seventh pin P 17  is set to a low level, so that the switching element Q 1  is turned off. 
     The fifth pint P 15  (ZCD) is a zero-cross detection terminal, and is connected to one terminal of a secondary coil n 2  of the inductor L 1  through a resistor R 13 . The other terminal of the secondary coil n 2  is connected to the circuit ground. Further, the fifth pin P 15  detects energy accumulated in the inductor L 1 , and if it is detected that the energy is no longer discharged from the inductor L 1 , the seventh pin P 17  is set to a high level, so that the switching element Q 1  is turned on. 
     The third pin P 13  (MULT) is an input terminal of an internal multiplier circuit (not shown), and detects the ripple voltage outputted from the rectifier circuit  3 . The ripple voltage is divided by a resistor R 5  and a series circuit including resistors R 2  to R 4 , and the divided voltage is inputted to the third pin P 13  of the PFC circuit IC 1 . Further, a capacitor C 7  is connected between the third pin P 13  and the circuit ground to remove the noise. 
     Further, the PFC circuit IC 1  controls such that the ON time of the switching element Q 1  gets longer as the ripple voltage increases and gets shorter as the ripple voltage decreases. Further, the internal multiplier circuit of the PFC circuit IC 1  connected to the third pin P 13  is used to control a peak value of the input current inputted from the commercial AC power source  200  through the full-wave rectifier DB 1  in a shape similar to a ripple voltage waveform. 
     The first pin P 11  (INV) is an inverting input terminal of an internal error amplifier, and the second pin P 12  (COMP) is an output terminal of the internal error amplifier. The first pin P 11  detects a DC voltage outputted from the step-up chopper circuit  4 . The DC voltage generated across the smoothing capacitor C 5  is divided by a series circuit including resistors R 6  to R 9  and a series circuit including a resistor R 10  and a variable resistor VR 1 , and the divided voltage is inputted to the first pin P 11 . Further, if the detection value is higher than a target voltage, it is controlled such that the ON time of the switching element Q 1  becomes shorter. If the detection value is lower than the target voltage, it is controlled such that the ON time of the switching element Q 1  becomes longer. Further, capacitors C 8  and C 9  and a resistor R 11  connected between the first pin P 11  and the second pin P 12  form a feedback impedance of the internal error amplifier of the PFC circuit IC 1 . 
     Next, the control power supply circuit  6  will be described. The control power supply circuit  6  of this embodiment includes an IPD element IC 2  and its peripheral circuits. The IPD element IC 2  is a so-called intelligence power device, and uses, e.g., MIP2E2D manufactured by Panasonic Corporation. 
     The IPD element IC 2  is a three-pin IC having a drain terminal P 21 , a source terminal P 22  and a control terminal P 23 . The IPD element  102  has a switching element including a power MOSFET and a control circuit for controlling a switching operation of the switching element. 
     Further, the internal switching element of the IPD element  102 , an inductor L 2 , a smoothing capacitor C 12  and a diode D 3  are included in a step-down chopper circuit. Specifically, the drain terminal P 21  of the IPD element IC 2  is connected to a positive electrode of the smoothing capacitor C 5 , and the source terminal P 22  is connected to a positive electrode of the smoothing capacitor C 12  through the inductor L 2 . Further, the diode D 3  is connected in parallel to a series circuit including the inductor L 2  and the smoothing capacitor C 12 , and a cathode of the diode D 3  is connected to the inductor L 2 . 
     Further, a power supply circuit of the IPD element IC 2  includes a Zener diode ZD 1 , a diode D 4 , a smoothing capacitor C 14 , and a capacitor C 15 . A parallel circuit including the smoothing capacitor C 14  and the capacitor C 15  is connected between the control terminal P 23  and the source terminal P 22  of the IPD element IC 2 . A positive electrode of the smoothing capacitor C 14  is connected to the control terminal P 23 . Further, a series circuit including the Zener diode ZD 1 , the diode D 4  and the smoothing capacitor C 14  is connected in parallel to the inductor L 2 . A cathode of the Zener diode ZD 1  is connected to the inductor L 2 , and a cathode of the diode D 4  is connected to the smoothing capacitor C 14 . Further, a capacitor C 13  is connected between the drain terminal P 21  of the IPD element  102  and the circuit ground to remove the noise. 
     In an initial stage when a power is inputted from the commercial AC power source  200 , the smoothing capacitor C 5  is charged by the ripple voltage outputted from the full-wave rectifier DB 1  through the inductor L 1  and the diode D 1 . Further, as the smoothing capacitor C 5  is charged, the current flows in a path including the drain terminal P 21  of the IPD element IC 2 →the control terminal P 23 →the smoothing capacitor C 14 →the inductor L 2 →the smoothing capacitor C 12 , thereby charging the smoothing capacitor C 14 . The voltage across the smoothing capacitor C 14  becomes an operation power supply to an internal control circuit of the IPD element  102 , so that the operation of the IPD element  102  is started and the switching operation of the internal switching element of the IPD element  102  is controlled. 
     If the switching element of the IPD element  102  is in an ON state, the current flows in a path including the smoothing capacitor C 5 →the drain terminal P 21 →the source terminal P 22 →the inductor L 2 →the smoothing capacitor C 12 , thereby charging the smoothing capacitor C 12 . Further, if the switching element of the IPD element  102  is in an OFF state, the accumulated energy at the inductor L 2  is discharged to the smoothing capacitor C 12  through the diode D 3 . By repeating the ON/OFF operation described above, the control voltage Vcc, to which the voltage across the smoothing capacitor C 5  is stepped down, is generated across the smoothing capacitor C 12 . 
     Further, if the switching element of the IPD element IC 2  is in an OFF state, a flyback current flows through the diode D 3 . However, in this case, the voltage across the inductor L 2  is clamped to the sum of the voltage across the smoothing capacitor C 12  and the forward voltage of the diode D 3 . The voltage obtained by subtracting the sum of the Zener voltage of the Zener diode ZD 1  and the forward voltage of the diode D 4  from the voltage across the inductor L 2  becomes the voltage across the smoothing capacitor C 14 . Further, the internal control circuit of the IPD element  102  controls the switching operation of the internal switching element of the IPD element  102  such that the voltage across the smoothing capacitor C 14  becomes constant. Accordingly, the voltage across the smoothing capacitor C 12  is controlled to be constant, and the smoothing capacitor C 14  is charged so that the IPD element  102  can be continuously driven. 
     The control power supply circuit  6  configured as described above supplies the control voltage Vcc to the step-up chopper controller  8 , the step-down converter controller  9  and the dimming controller  10  while the voltage across the smoothing capacitor C 12  serves as the output voltage thereof. Hereinafter, a portion having the same potential as the control voltage Vcc is referred to as a control power supply. 
     Next, the step-down converter  5  for stepping down the DC voltage generated across the smoothing capacitor C 5  will be described. 
     The step-down converter  5  includes a step-down chopper circuit including a switching element Q 2 , an inductor L 3 , a smoothing capacitor C 16 , and a diode D 5 . Specifically, a series circuit including the switching element Q 2 , the inductor L 3  and the smoothing capacitor C 16  is connected in parallel to the smoothing capacitor C 5 . The diode D 5  is connected in parallel to the series circuit of the inductor L 3  and the smoothing capacitor C 16 . The switching element Q 2  includes an n channel MOSFET, and has a drain terminal connected to a positive electrode of the smoothing capacitor C 5 , and a source terminal connected to a positive electrode of the smoothing capacitor C 16  through the inductor L 3 . Further, an anode of the diode D 5  is connected to a negative electrode of the smoothing capacitor C 16  and a cathode of the diode D 5  is connected to the inductor L 3 . 
     Further, if the switching element Q 2  is turned on, the current flows, from the smoothing capacitor C 5 , in a path including the switching element Q 2 →the inductor L 3  the smoothing capacitor C 16 . Further, if the switching element Q 2  is turned off, the energy accumulated in the inductor L 3  is discharged to the smoothing capacitor C 16  through the diode D 5 . Further, by repeating the ON/OFF operation described above, the voltage, to which the DC voltage across the smoothing capacitor C 5  is stepped down, is generated across the smoothing capacitor C 16 . 
     The step-down converter  5  configured as described above controls the current supplied to a load  12  (hereinafter, referred to as LED current Io) to be constant while the voltage across the smoothing capacitor C 16  serves as the output voltage thereof. The load  12  is configured by connecting LED modules  122  in parallel, each LED module having LED elements  121  connected in series to each other. The load  12  of this embodiment is configured by connecting two LED modules  122  in parallel. The LED modules  122  may be respectively referred to as LED modules  122   a  and  122   b . Further, the current detector  7  is connected in series to the LED module  122   a . Further, each of the LED elements  121  is turned on by the LED current Io supplied from the step-down converter  5 . 
     Next, the step-down converter controller  9  will be described. 
     The step-down converter controller  9  includes timer integrated circuits IC 3  and IC 4 , and their peripheral circuits. The timer integrated circuits IC 3  and IC 4  are well-known timer ICs (so-called 555 timer circuits), and may employ, e.g., μPD5555 manufactured by Renesas Electronics Corporation, μPD5556 of its dual version, or a compatible product thereof. 
     The timer integrated circuits IC 3  and IC 4  include first pins P 31  and P 41  to eighth pins P 38  and P 48 , respectively, to which the peripheral circuits are connected. Hereinafter, the function and operation of the first pins P 31  and P 41  to the eighth pins P 38  and P 48  of the timer integrated circuits IC 3  and IC 4  will be described. 
     The eighth pins P 38  and P 48  are power supply terminals and the first pins P 31  and P 41  are ground terminals. The control voltage Vcc is supplied between each of the eighth pins P 38  and P 48  and the corresponding first pins P 31  and P 41 . Further, a capacitor C 17  is connected between the eighth pin P 38  and the first pin P 31  of the timer integrated circuit IC 3 . A capacitor C 18  is connected between the eighth pin P 48  and the first pin P 41  of the timer integrated circuit IC 4 . The capacitors C 17  and C 18  are small-capacity capacitors for power supply bypass to remove the noise of the control voltage Vcc. 
     The fifth pins P 35  and P 45  are control terminals, and a reference voltage Vb 1  that is ⅔ of the control voltage Vcc is applied to each of the fifth pins P 35  and P 45  by an internal resistor divider. Further, a capacitor C 19  is connected between the fifth pin P 35  and the first pin P 31  of the timer integrated circuit IC 3 . A capacitor C 20  is connected between the fifth pin P 45  and the first pin P 41  of the timer integrated circuit IC 4 . The capacitors C 19  and C 20  are small-capacity capacitors for bypass to remove the noise of the reference voltage Vb 1  applied to each of the fifth pins P 35  and P 45 . 
     The sixth pins P 36  and P 46  are threshold terminals, and if a voltage applied to each of the sixth pins P 36  and P 46  is higher than the reference voltage Vb 1 , an internal flip-flop is inverted. 
     Further, the output level of each of the third pins P 33  and P 43  serving as output terminals becomes a low level. Further, the seventh pins P 37  and P 47  serving as discharge terminals are short-circuited to the first pins P 31  and P 41  (circuit ground), respectively. 
     The second pins P 32  and P 42  are trigger terminals, and if a voltage applied to each of the second pins  932  and  942  is lower than a reference voltage Vb 2  that is ½ of the reference voltage Vb 1 , an internal flip-flop is inverted. Further, the output level of each of the third pins  933  and  943  becomes a high level, and the seventh pins  937  and P 47  turn into an open-circuit state. 
     The fourth pins P 34  and P 44  are reset terminals. If a voltage applied to each of the fourth pins P 34  and P 44  is less than 2V, the operation is stopped and the output level of each of the third pins P 33  and P 43  is fixed to a low level. 
     Next, an operation of each of the timer integrated circuits IC 3  and IC 4  will be described in detail. Hereinafter, the timer integrated circuit IC 3  is referred to as a high frequency oscillation circuit IC 3 , and the timer integrated circuit IC 4  is referred to as a pulse width setting circuit IC 4 . 
     First, an operation of the high frequency oscillation circuit IC 3  will be described in detail. 
     Resistors R 17  and R 18  and a capacitor C 21  which determine a time constant are connected, as peripheral circuits, to the high frequency oscillation circuit IC 3 , and the high frequency oscillation circuit IC 3  operates as an astable multivibrator. 
     A series circuit including the resistors R 17  and R 18  and the capacitor C 21  is connected between the control power supply and the circuit ground. A connection node between the resistors R 17  and R 18  is connected to the seventh pin P 37 , and a connection node between the resistor R 18  and the capacitor C 21  is connected to the second pin P 32  and the sixth pin P 36 . 
     Further, the voltage across the capacitor C 21  is applied to the second pin P 32  and the sixth pin P 36  to be compared with the reference voltages Vb 2  and Vb 1 , respectively. 
     In an initial power input, since the voltage across the capacitor C 21  is lower than the reference voltage Vb 2  at the second pin P 32 , the output level of the third pin P 33  becomes a high level, and the seventh pin P 37  is in an open-circuit state. Accordingly, the current flows through the capacitor C 21  from the control power supply through the resistors R 17  and R 18 , thereby charging the capacitor C 21 . 
     By the charging operation, if the capacitor C 21  is charged and the voltage across the capacitor C 21  becomes higher than the reference voltage Vb 1  at the sixth pin P 36 , the output level of the third pin P 33  becomes a low level and the seventh pin P 37  is short-circuited to the first pin P 31 . Accordingly, the current flows from the capacitor C 21  to the circuit ground through the resistor R 18 , thereby discharging the capacitor C 21 . 
     By the discharging operation, the capacitor C 21  is discharged, and the voltage across the capacitor C 21  decreases. If the voltage across the capacitor C 21  is lower than the reference voltage Vb 2  at the second pin P 32 , the output level of the third pin P 33  becomes a high level, and the seventh pin P 37  turns into an open-circuit state. Accordingly, the capacitor C 21  gets charged again. Then, the above-described charging operation and discharging operation are repeatedly performed. 
     The time constant determined by the resistors R 17  and R 18  and the capacitor C 21  is set such that the oscillation frequency of the third pin P 33  is several tens of kHz. 
     Further, the resistance of the resistor R 17  is set to be sufficiently smaller than the resistance of the resistor R 18 . Thus, the period during which the capacitor C 21  has been charged (the third pin P 33  has a low level) is extremely reduced. Accordingly, at the third pin P 33 , a pulse signal having a short low level pulse width is repeatedly outputted at a frequency of several tens of kHz. The second pin P 42  of the pulse width setting circuit IC 4  is triggered only once every cycle by using a falling edge of the pulse signal. 
     Next, an operation of the pulse width setting circuit IC 4  will be described in detail. 
     The resistor R 19  and a variable resistor VR 2  and a capacitor C 22  which determine a time constant are connected, as peripheral circuits, to the pulse width setting circuit IC 4 , and the pulse width setting circuit IC 4  operates as a monostable multivibrator. A series circuit including the variable resistor VR 2  and the resistor R 19  and the capacitor C 22  is connected between the control power supply and the circuit ground. The sixth pin P 46  and the seventh pin P 47  are connected to a connection node between the resistor R 19  and the capacitor C 22 . Further, a light receiving element PC 11  of a photocoupler PC 1  is connected in parallel to a series circuit including the R 19  and the variable resistor VR 2 . The pulse width of the monostable multivibrator is variably controlled based on the intensity of an optical signal of a light emitting element PC 12  of the photocoupler PC 1 . 
     The second pin P 42  of the pulse width setting circuit IC 4  is connected to the third pin P 33  of the high frequency oscillation circuit IC 3  and a pulse signal having a short low level pulse width is inputted thereto from the third pin P 33  of the high frequency oscillation circuit IC 3 . Further, at a falling edge of the pulse signal, the third pin P 43  of the pulse width setting circuit IC 4  has a high level and the seventh pin P 47  is in an open state. Accordingly, the capacitor C 22  is charged by the control power supply through the series circuit including the resistor R 19  and the variable resistor VR 2  and the light receiving element PC 11  of the photocoupler PC 1 . 
     If the voltage across the capacitor C 22  becomes higher than the reference voltage Vb 1  at the sixth pin P 46  by the charging operation, the output level of the third pin P 43  becomes a low level, and the seventh pin P 47  becomes short-circuited to the first pin P 41 . Accordingly, the capacitor C 22  is discharged instantaneously. 
     Accordingly, the high level period of the pulse signal outputted from the third pin P 43  of the pulse width setting circuit IC 4  is determined by the time required for charging the capacitor C 22  from the ground potential to the reference voltage Vb 2 . The maximum value of the charging time is set to be shorter than the oscillation period of the high frequency oscillation circuit IC 3 . Further, the minimum value of the charging time is set to be longer than the low level period of the pulse signal outputted from the third pin P 33  of the high frequency oscillation circuit IC 3 . 
     The third pin P 43  is connected to a parallel circuit including an electrolytic capacitor C 23  and a diode D 6  through a primary coil T 11  of the transformer T 1 . 
     One terminal of the primary coil T 11  of the transformer T 1  is connected to the third pin P 43 , and the other terminal of the primary coil T 11  is connected to a positive electrode of the electrolytic capacitor C 23  and a cathode of the diode D 6 . Further, a series circuit including resistors R 20  and R 21  is connected between both terminals of a secondary coil T 12  of the transformer T 1 . One terminal of the secondary coil T 12  is connected to the source of the switching element Q 2 . Further, the resistor R 21  is connected between the source and gate of the switching element Q 2 . Further, a series circuit including a diode D 7  and a resistor R 22  is connected in parallel to the resistor R 20 . An anode of the diode D 7  is connected to the gate of the switching element Q 2 . 
     Further, a switching operation of the switching element Q 2  is controlled by using the pulse signal outputted from the third pin P 43  of the pulse width setting circuit IC 4 . 
     If the pulse signal outputted from the third pin P 43  is of a high level, the current flows in the electrolytic capacitor C 23  through the primary coil T 11  of the transformer T 1  to thereby charge the electrolytic capacitor C 23 . 
     In this case, an induced electromotive force is generated at the secondary coil T 12  of the transformer T 1 , and the current flows through the resistors R 20  and R 21 , so that the voltage across the resistor R 21  increases. Further, if the voltage across the resistor R 21  becomes equal to or higher than the gate-source threshold voltage of the switching element Q 2 , the switching element Q 2  is turned on. 
     Further, if the pulse signal outputted from the third pin P 43  is of a low level, the current flows from the electrolytic capacitor C 23  through the primary coil T 11 . Accordingly, at the secondary coil T 12 , the electric charges between the gate and source of the switching element Q 2  are discharged through the diode D 7  and the resistor R 22 , so that the switching element Q 2  is turned off. 
     By repeating the above operation, the pulse width setting circuit IC 4  controls the switching operation of the switching element Q 2 . 
     Further, the control voltage Vcc is applied to the fourth pin P 34  of the high frequency oscillation circuit IC 3 , and a voltage obtained by dividing the control voltage Vcc by resistors R 23  and R 24  is applied to the fourth pin P 44  of the pulse width setting circuit IC 4 . Accordingly, after the control power supply circuit  6  is driven to output the control voltage Vcc, the high frequency oscillation circuit IC 3  and the pulse width setting circuit IC 4  are driven. 
     Next, the dimming controller  10  will be described. 
     A dimming signal inputted to the dimming controller  10  is a PWM signal including a square wave voltage signal with a variable pulse width, having a frequency of 1 kHz and an amplitude of 10 V. The dimming signal is widely used as a dimming signal of an inverter lighting device of a fluorescent lamp. Further, a dimming signal line through which the dimming signal is transmitted is provided in each illumination apparatus separately from a power line. 
     A full-wave rectifier DB 2  is connected to an input terminal of the dimming controller  10  of this embodiment. Accordingly, even though a dimming signal line is connected with reverse polarity, the dimming controller  10  operates normally. A series circuit including resistors R 25  and R 26  and a light emitting element PC 22  of a photocoupler PC 2  is connected to an output terminal of the full-wave rectifier DB 2 . A Zener diode ZD 2  is connected in parallel to a series circuit including the resistor R 26  and the light emitting element PC 22 . 
     The photocoupler PC 2  functions as an insulation circuit. Generally, a plurality of illumination apparatuses is connected in parallel to the dimming signal line and the power line. In such case, since the circuit ground of each illumination apparatus does not have the same potential, it is necessary to insulate the dimming signal line from the circuit ground of each illumination apparatus. 
     The light emitting element PC 22  of the photocoupler PC 2  is connected to the dimming signal line through the resistors R 25  and R 26  and the full-wave rectifier DB 2 . Further, a series circuit including a light receiving element PC 21  of the photocoupler PC 2  and a resistor R 27  is connected between the control power supply and the circuit ground. 
     If the dimming signal (PWM signal) inputted through the dimming signal line is of a high level, the luminous flux from the light emitting element PC 22  of the photocoupler PC 2  increases, so that the on-resistance of the light receiving element PC 21  decreases and the current flowing through the light receiving element PC 21  increases. Accordingly, the voltage at a connection node between the resistor R 27  and the light receiving element PC 21  decreases. Hereinafter, the voltage at the connection node between the resistor R 27  and the light receiving element PC 21  is referred to as a dimming voltage. 
     Further, if the dimming signal is of a low level, the luminous flux from the light emitting element PC 22  decreases, so that the on-resistance of the light receiving element PC 21  increases and the current flowing in the light receiving element PC 21  decreases. Accordingly, the dimming voltage increases. 
     The dimming voltage is inputted to an integrated circuit IC 5  (hereinafter, referred to as a dimming circuit IC 5 ) including operational amplifiers A 1  and A 2 . The dimming circuit IC 5 , a resistor R 28  and a capacitor C 24  are included in a DC conversion circuit. A change in the dimming voltage is repeated at a frequency (1 kHz) of the dimming signal, but is smoothed by a time constant circuit including the resistor R 28  and the capacitor C 24  to be converted into a DC voltage. 
     The dimming circuit IC 5  employs, e.g., μPC358 manufactured by Renesas Electronics Corporation or a compatible product thereof. The dimming circuit IC 5  is driven by the supply of the control voltage Vcc. 
     The operational amplifier A 1  is used as a buffer amplifier. In the operational amplifier A 1 , a dimming voltage is applied to a non-inverting input terminal, an inverting input terminal is connected to an output terminal, and the output terminal is connected to the circuit ground through a series circuit including the resistor R 28  and the smoothing capacitor C 24 . Further, the operational amplifier A 1  converts the high impedance input dimming voltage into a low impedance output voltage, and performs charging and discharging of the smoothing capacitor C 24  through the resistor R 28 . 
     If a low level period of the dimming signal is long, the period during which the capacitor C 24  is charged through the resistor R 28  becomes long, so that the voltage across the smoothing capacitor C 24  increases. Further, if a high level period of the dimming signal is long, the period during which the capacitor C 24  is discharged through the resistor R 28  becomes long, so that the voltage across the smoothing capacitor C 24  decreases. 
     The operational amplifier A 2  is used as a buffer amplifier, and a positive electrode of the smoothing capacitor C 24  is connected to a non-inverting input terminal thereof. Further, an inverting input terminal of the operational amplifier A 2  is connected to an output terminal of the operational amplifier A 2 , and the output terminal is connected to the control power supply through the light emitting element PC 12  of the photocoupler PC 1  and the resistor R 29 . Further, the high impedance input voltage across the capacitor C 24  is converted into a low impedance output voltage by the buffer amplifier including the operational amplifier A 2  and, then, the low impedance voltage is outputted, so that the light emitting element PC 12  of the photocoupler PC 1  is driven. 
     When the voltage across the smoothing capacitor C 24  is low, the output voltage of the operational amplifier A 2  is also low. Accordingly, the current flowing in the light emitting element PC 12  from the control power supply through the resistor R 29  increases, so that the luminous flux increases. Consequently, the on-resistance of the light receiving element PC 11  decreases, and the current flowing in the light receiving element PC 11  increases. That is, if the high level period of the dimming signal becomes long, the ON pulse width of the switching element Q 2  set by the pulse width setting circuit IC 4  is reduced, so that the LED current Io outputted from the step-down converter  5  decreases. 
     Further, if the voltage across the smoothing capacitor C 24  is high, the output voltage of the operational amplifier A 2  becomes high. Accordingly, the current flowing in the light emitting element PC 12  from the control power supply through the resistor R 29  decreases, so that the luminous flux decreases. Consequently, the on-resistance of the light receiving element PC 11  increases, and the current flowing in the light receiving element PC 11  decreases. That is, if the low level period of the dimming signal becomes long, the ON pulse width of the switching element Q 2  set by the pulse width setting circuit IC 4  becomes long, so that the LED current Io outputted from the step-down converter  5  increases. 
     Further, in a case where the dimming signal line is disconnected, the dimming signal always becomes to be of a low level, so that the LED current Io becomes to be of a maximum level and all lights are turned on. 
     Further, the step-down converter  5 , the step-down converter controller  9  and the dimming controller  10  correspond to a constant current supply unit described in the claims. Further, the filter circuit  2 , the rectifier circuit  3 , the step-up chopper circuit  4 , the step-down converter  5 , the control power supply circuit  6 , the step-up chopper controller  8 , the step-down converter controller  9 , and the dimming controller  10  correspond to a lighting unit described in the claims. 
     Next, the current detector  7  and the abnormality detector  11  will be described with reference to  FIG. 3 . 
     The current detector  7  is configured as a resistor R 30 , and connected in series to the LED module  122   a  to detect the current flowing through the LED module  122   a.    
     The abnormality detector  11  detects an abnormality in the load  12  based on an increase/decrease of a voltage across the resistor R 30 . The abnormality detector  11  includes switching elements Q 3  to Q 5 , resistors R 31  to R 35 , a comparator CP 1 , and a reference voltage generator E 1 . Although the current detector  7  is included in the abnormality detector  11  in  FIG. 3  for convenience of illustration, the current detector  7  is not a component of the abnormality detector  11 . 
     A series circuit including the resistor R 31  and the switching element Q 3  is connected between outputs (between the control power supply and the circuit ground) of the control power supply circuit  6 . The switching element Q 3  includes an NPN transistor having a collector connected to the control power supply through the resistor R 31  and an emitter connected to the circuit ground. Further, a series circuit including the resistors R 30  and R 32  is connected between a base and the emitter of the switching element Q 3 . A voltage across the resistor R 30  is applied to the base of the switching element Q 3  through the resistor R 32 . 
     Further, the collector of the switching element Q 3  is connected to a resistor R 33  and the switching element Q 4 . The switching element Q 4  includes an NPN transistor having an emitter connected to the circuit ground. The resistor R 33  is connected between a base and the emitter of the switching element Q 4 , and a voltage across the resistor R 30  is applied to the base of the switching element Q 4 . 
     Further, a non-inverting input terminal of the comparator CP 1  is connected to the resistor R 30  through the resistor R 34 , and the voltage across the resistor R 30  is applied to the non-inverting input terminal. Further, an inverting input terminal of the comparator CP 1  is connected to the reference voltage generator E 1 , and a reference voltage Vb 3  is applied to the inverting input terminal. An output terminal of the comparator CP 1  is connected to a base of the switching element Q 5  including an NPN transistor through the resistor R 35 . Further, an emitter of the switching element Q 5  is connected to the circuit ground. 
     Further, the abnormality detector  11  detects an abnormality in the load  12  based on whether the voltage across the resistor R 30  is within a predetermined range. If the voltage across the resistor R 30  is within the predetermined range, the abnormality detector  11  has an output state in which the abnormality in the load  12  is not detected. If the voltage across the resistor R 30  is out of the predetermined range, the abnormality detector  11  has an output state in which the abnormality in the load  12  is detected. In other words, if the current flowing in the LED module  122   a  is larger than an upper limit, or smaller than a lower limit of the predetermined current range, it is determined that an abnormality in the load  12  is detected. Further, if the abnormality detector  11  detects the abnormality in the load  12 , the output state thereof is switched by turning on either the switching element Q 4  or the switching element Q 5  based on that the current flowing in the LED module  122   a  is larger than an upper limit, or smaller than a lower limit. 
     For example, if the LED module  122   a  is detached or in an open-circuit mode failure, or if the LED module  122   b  is in a short-circuit mode failure, the current does not flow through the LED module  122   a . Accordingly, the voltage across the resistor R 30  is reduced to almost zero, and the switching element Q 3  is turned off. When the switching element Q 3  is turned off, the voltage across the resistor R 33  increases and the switching element Q 4  is turned on. Further, the open-circuit mode failure indicates a failure in a state where both terminals of the LED module  122  are insulated, and the short-circuit mode failure indicates a failure in a state where both terminals of the LED module  122  are short-circuited. 
     Further, if the LED module  122   b  is detached or in an open-circuit mode failure, or if the LED module  122   a  is in a short-circuit mode failure, the current flowing through the LED module  122   a  increases. Accordingly, the voltage across the resistor R 30  increases. If the voltage across the resistor R 30  is higher than the reference voltage Vb 3 , the output level of the comparator CP 1  becomes a high level, and the switching element Q 5  is turned on. 
     That is, if the voltage across the resistor R 30  is equal to or higher than an upper limit of a predetermined range, the switching element Q 5  is turned on. If the voltage across the resistor R 30  is equal to or lower than a lower limit of the predetermined range, the switching element Q 4  is turned on. 
     Further, each collector of the switching elements Q 4  and Q 5  is connected to at least one of the fourth pin P 44  of the pulse width setting circuit IC 4 , the fifth pint P 15  of the PFC circuit IC 1 , and the non-inverting input terminal of the operational amplifier A 2  of the dimming circuit IC 5 . 
     In a case where the collectors of the switching elements Q 4  and Q 5  are connected to the fourth pin P 44  of the pulse width setting circuit IC 4 , for example, if one of the switching elements Q 4  and Q 5  is turned on, the fourth pin P 44  is short-circuited to the circuit ground. Accordingly, since the operation of the pulse width setting circuit IC 4  is stopped, and the switching operation of the switching element Q 2  is stopped, the LED current Io is not supplied to the load  12 . 
     In a case where the collectors of the switching elements Q 4  and Q 5  are connected to the fifth pin P 15  of the PFC circuit IC 1 , for example, if one of the switching elements Q 4  and Q 5  is turned on, the fifth pin P 15  is short-circuited to the circuit ground. Accordingly, since the operation of the switching element Q 1  is stopped, the LED current Io is not supplied to the load  12 . 
     In a case where the collectors of the switching elements Q 4  and Q 5  are connected to the non-inverting input terminal of the operational amplifier A 2  of the dimming circuit IC 5 , for example, if one of the switching elements Q 4  and Q 5  is turned on, the positive electrode of the capacitor C 24  is short-circuited to the circuit ground. Accordingly, the ON pulse width of the switching element Q 2  decreases, and the LED current Io is reduced (suppressed). 
     Further, it may be configured to increase the voltage applied to the first pin P 11  of the PFC circuit IC 1  by turning on one of the switching elements Q 4  and Q 5 . Accordingly, the output of the step-up chopper circuit  4  is suppressed and, thus, the LED current Io is reduced (suppressed). 
     Further, the collectors of the switching elements Q 4  and Q 5  may be connected to the same location or different locations of the above-mentioned locations. Further, the collectors of the switching elements Q 4  and Q 5  may be connected to multiple locations of the above-mentioned locations. 
     Thus, in this embodiment, the current flowing in only one LED module  122  among the LED modules  122  connected in parallel to each other is detected, and the presence of an abnormality in the load  12  is detected based on the detected current value. Further, if the abnormality in the load  12  is detected, the LED current Io is reduced, thereby preventing the concentrated current from flowing through the normally operating LED module  122 . 
     Further, since there is no need to provide an abnormality detection unit for each of the LED modules  122 , the circuit configuration becomes simple, thereby reducing the costs. Further, since the current detector  7  is provided only for one module, i.e., the LED module  122   a , the power loss due to the current detector  7  is suppressed, and the overall conversion efficiency of the lighting device  1  is improved. 
     Further, in this embodiment, the constant current supply unit (the step-down converter  5 , the step-down converter controller  9  and the dimming controller  10 ) is used and the LED current Io (constant current) is commonly supplied to the LED modules  122 . Accordingly, since the constant current circuit is not provided for each of the LED modules  122 , the power loss due to the constant current circuit is suppressed, and the overall conversion efficiency of the lighting device  1  is improved. 
     Further, in this embodiment, the collectors of the switching elements Q 4  and Q 5  are connected to the non-inverting input terminal of the operational amplifier A 2  of the dimming circuit IC 5 , and if the abnormality in the load  12  is detected, the LED current Io is reduced. Accordingly, the normally operating LED modules  122  may be continuously turned on. 
     Further, the number of the LED modules  122  is not limited to two, and three or more LED modules  122  may be included in the load  12 . For example, as shown in  FIG. 4 , five LED modules  122   a  to  122   e  may form the load  12 . Also in this case, if any one of the LED modules  122   b  to  122   e  other than the LED module  122   a  is detached or in an open-circuit mode failure, or if the LED module  122   a  is in a short-circuit mode failure, the current flowing through the LED module  122   a  increases. Further, if the LED module  122   a  is detached or in an open-circuit mode failure, or if the LED modules  122   b  to  122   e  other than the LED module  122   a  are in a short-circuit mode failure, the current flowing through the LED module  122   a  decreases. Accordingly, the LED lighting device  1  can detect the presence of an abnormality in the entire load  12 . 
     Further, the configuration of the abnormality detector  11  is not limited thereto. For example, as shown in  FIG. 4 , it may be configured such that resistors R 36  and R 37  and a switching element Q 6  are included in an abnormality detector lie so as to detect an increase in the voltage across the resistor R 30 . The switching element Q 6  has a collector connected to the control power supply, an emitter connected to the circuit ground through the resistor R 37 , and a base connected to the resistor R 30  through the resistor R 36 . Further, the emitter of the switching element Q 6  is connected to the first pin P 11  of the PFC circuit IC 1 . The PFC circuit IC 1  detects the presence of an abnormality in the load  12  based on the detected value of the abnormality detector  11   a . If the abnormality in the load  12  is detected, the LED current Io is reduced (suppressed). Further, in this case, the abnormality detector  11   a  and the PFC circuit IC 1  correspond to an abnormality detection unit described in the claims. 
     Specifically, as the number of the LED modules  122  which are detached or in an open-circuit mode failure among the LED modules  122   b  to  122   e  increases, the voltage across the resistor R 30  continuously increases. Accordingly, the on-resistance of the switching element Q 6  decreases, and the current flowing between the collector and emitter continuously increases. Further, since the voltage across the resistor R 37  continuously increases, the voltage applied to the first pin P 11  of the PFC circuit IC 1  also continuously increases. Accordingly, the output of the step-up chopper circuit  4  is continuously reduced, and the LED current Io also is continuously reduced. 
     That is, as the number of the LED modules  122  which are detached or in an open-circuit mode failure among the LED modules  122   b  to  122   e  other than the LED module  122   a  increases, a difference between the current value flowing through the LED module  122   a  and the upper limit of the current range increases. Therefore, as the difference increases, the lighting device  1  of this embodiment increases a reduction in the LED current Io to thereby prevent the excessive current from flowing through the normally operating LED modules  122 . 
     Further, it may be configured such that as a difference between the current value flowing through the LED module  122   a  and the lower limit of the current range increases, a reduction in the LED current Io increases. Accordingly, it is possible to prevent the excessive current from flowing through the normally operating LED modules  122 . 
     Further, the circuit configuration of the step-down converter  5  of this embodiment includes the switching element Q 2 , the diode D 5 , the inductor L 3  and the smoothing capacitor C 16  as shown in  FIG. 2 , but it is not limited thereto. 
     For example, a step-up chopper circuit  51  shown in  FIG. 5A  may be employed instead. The step-up chopper circuit  51  includes a series circuit of an inductor L 3   a  and a switching element Q 2   a , and a series circuit of a diode D 5   a  and a smoothing capacitor C 16   a  connected in parallel to a switching element Q 2   a.    
     Further, a step-up/down chopper circuit  52  shown in  FIG. 5B  may be employed instead. The step-up/down chopper circuit  52  includes a series circuit of an inductor L 3   b ) and a switching element Q 2   b , and a series circuit of a diode D 5   b  and a smoothing capacitor C 16   b  connected in parallel to the inductor L 3   b.    
     Further, a flyback converter circuit  53  shown in  FIG. 5C  may be employed instead. The flyback converter circuit  53  includes a switching element Q 2   c  connected to a primary coil T 21   c  of a transformer T 2   c , and a series circuit of a diode D 5   c  and a smoothing capacitor C 16   c  connected to both terminals of a secondary coil T 22   c . Further, the primary coil T 21   c  and the secondary coil T 22   c  of the transformer T 2   c  have the same polarity. 
     Further, a fly-forward converter circuit  54  shown in  FIG. 5D  may be employed instead. The fly-forward converter circuit  54  includes a switching element Q 2   d  connected to a primary coil T 21   d  of a transformer T 2   d , and a series circuit of a diode D 5   d  and a smoothing capacitor C 16   d  connected to both terminals of a secondary coil T 22   d . Further, the primary coil T 21   d  and the secondary coil T 22   d  of the transformer T 2   d  have the opposite polarity. 
     Further, as shown in  FIG. 5E , a step-down converter circuit  55  having a switching element Q 2   e  provided on a low side may be employed instead. The step-down converter circuit  55  includes a series circuit of a diode D 5   e  and a switching element Q 2   e , and a series circuit of an inductor L 3   e  and a smoothing capacitor C 16   e  connected in parallel to the diode D 5   e.    
     Further, the circuit configuration of the step-up chopper circuit  4  of this embodiment includes, as shown in  FIG. 2 , the inductor L 1 , the switching element Q 1 , the diode D 1  and the smoothing capacitor C 5 , but it is not limited thereto. 
     For example, the flyback converter circuit  53  shown in  FIG. 5C  may be employed thereto. 
     Further, in this embodiment, the LED elements  121  are used as semiconductor light emitting elements, but it is not limited thereto. For example, organic EL elements or semiconductor laser elements may be used as semiconductor light emitting elements. 
     Second Embodiment 
       FIG. 6  illustrates a block diagram of the lighting device  1  in accordance with a second embodiment of the present invention. The lighting device  1  of this embodiment includes a timer circuit  13 . The like reference numerals will be given to the like parts as those in the first embodiment, and redundant description thereof will be omitted. Further, in this embodiment, the filter circuit  2 , the rectifier circuit  3 , the step-up chopper circuit  4 , the step-down converter  5 , the control power supply circuit  6 , the step-up chopper controller  8 , the step-down converter controller  9 , the dimming controller  10  and the timer circuit  13  correspond to a lighting unit described in the claims. 
     The timer circuit  13  alternately and repeatedly blocks and unblocks the output of the abnormality detector  11  when it is determined that the load  12  is in an abnormal state based on the output of the abnormality detector  11 . For example, a case where the abnormality detector  11  is configured as shown in  FIG. 3  and the collectors of the switching elements Q 4  and Q 5  are connected to the fourth pin P 44  of the pulse width setting circuit IC 4  will be described. In this case, when the abnormality detector  11  detects an abnormality in the load  12 , the timer circuit  13  alternately and repeatedly blocks and unblocks electric conduction between the fourth pin P 44  and the collectors of the switching elements Q 4  and Q 5 . 
     If the timer circuit  13  allows electric conduction between the fourth pin P 44  and the collectors of the switching elements Q 4  and Q 5 , since the fourth pin P 44  of the pulse width setting circuit IC 4  is short-circuited to the circuit ground, the supply of the LED current Io is stopped. Further, if the timer circuit  13  blocks electric conduction between the fourth pin P 44  and the collectors of the switching elements Q 4  and Q 5 , the LED current Io in a normal state is supplied to the load  12  even though the load  12  is in an abnormal state. 
     That is, the timer circuit  13  performs an intermittent operation for intermittently reducing the LED current Io when it is determined that the load  12  is in an abnormal state based on the output of the abnormality detector  11 . Accordingly, the LED current Io supplied to the load  12  is suppressed, and it is possible to prevent the concentrated current from flowing through the normally operating LED modules  122 , and further to continuously turn on the normally operating LED modules  122 . 
     Further, if the abnormality in the load  12  is eliminated and the load  12  returns to a normal state due to replacement or reinstallation of the LED modules  122  while the timer circuit  13  repeatedly performs the conduction blocking operation, the timer circuit  13  stops the conduction blocking operation. Accordingly, the LED current Io flowing in a normal state is supplied from the step-down converter  5  to the load  12 , thereby normally turning on the load  12 . That is, if the abnormality detector  11  is switched from a state in which an abnormality in the load  12  is detected to a state in which the abnormality in the load  12  is not detected while the timer circuit  13  performs an intermittent operation, the timer circuit  13  stops the intermittent operation. In this embodiment, when the abnormality in the load  12  is eliminated, the load  12  can be automatically restored to the ON state. 
     Further, a case where the collectors of the switching elements Q 4  and Q 5  are connected to the fourth pin P 44  of the pulse width setting circuit IC 4  has been described in this embodiment, but it is not limited thereto. In the similar way as the first embodiment, even when the collectors of the switching elements Q 4  and Q 5  are connected to the fifth pin P 15  of the PFC circuit IC 1 , the same effect can be obtained. 
     Further, the collectors of the switching elements Q 4  and Q 5  may be connected to the non-inverting input terminal of the operational amplifier A 2  of the dimming circuit IC 5 . 
     Further, the abnormality detector  11  may be configured as an abnormality detector  11   a  shown in  FIG. 4 . 
     Third Embodiment 
       FIG. 7  illustrates an external appearance of an illumination apparatus in accordance with a third embodiment of the present invention. In this illumination apparatus, the lighting device  1  is separately provided from an LED unit  14 . 
     The LED unit  14  is configured such that a substrate  142  in which the load  12  having a plurality of the LED modules  122  is mounted is contained in a metal cylindrical housing  141  having an open side, and the open side of the housing  141  is covered with a light diffusion plate  143 . The light emitted from the LED modules  122  is irradiated to the outside after being diffused and transmitted through the light diffusion plate  143 . The LED unit  14  is embedded in a ceiling panel  15  such that the light diffusion plate  143  is exposed downward from the surface of the ceiling panel  15 . 
     The lighting device  1  is disposed on the rear surface of the ceiling panel  15 . The step-down converter  5  is connected to the LED unit  14  through a lead line  16  and a connector  17  such that the LED current Io is supplied to the LED unit  14 . The connector  17  is configured such that a connector  171  on the side of the lighting device  1  is detachably attachable to a connector  172  on the side of the LED unit  14 . The lighting device  1  and the LED unit  14  can be separated from each other during maintenance or the like. 
     The lighting device  1  has a circuit configuration same as those of the first and second embodiments. Therefore, in the illumination apparatus described above, the LED current Io is reduced when an abnormality in the load  12  is detected in the LED unit  14 . 
     Further, the lighting device  1  and the LED unit  14  may be contained in the same housing. 
     Further, the lighting device  1  may be used to turn on a backlight of an LCD monitor, a light source of a copying machine, scanner or projector or the like as well as being used in the illumination apparatus. 
     While the invention has been shown and described with respect to the embodiments, it will be understood by those skilled in the art that various changes and modification may be made without departing from the scope of the invention as defined in the following claims.