Patent Publication Number: US-11651736-B2

Title: Electronic display with hybrid in-pixel and external compensation

Description:
This application is a continuation of patent application Ser. No. 17/062,786, filed Oct. 5, 2020, which is a continuation of patent application Ser. No. 16/716,911, filed Dec. 17, 2019, now U.S. Pat. No. 10,916,198, which claims the benefit of provisional patent application No. 62/791,522, filed Jan. 11, 2019, all of which are hereby incorporated by reference herein in their entireties. 
    
    
     BACKGROUND 
     This relates generally to electronic devices with displays and, more particularly, to display driver circuitry for displays such as organic-light-emitting diode displays. 
     Electronic devices often include displays. For example, cellular telephones and portable computers include displays for presenting information to users. 
     Displays such as organic light-emitting diode displays have an array of display pixels based on light-emitting diodes. In this type of display, each display pixel includes a light-emitting diode and thin-film transistors for controlling application of a signal to the light-emitting diode to produce light. 
     An organic light-emitting diode display pixel includes a drive thin-film transistor connected to a data line via an access thin-film transistor. The access transistor may have a gate terminal that receives a scan signal via a corresponding scan line. Image data on the data line can be loaded into the display pixel by asserting the scan signal to turn on the access transistor. The display pixel further includes a current source transistor that provides current to the organic light-emitting diode to produce light. 
     Transistors in an organic light-emitting diode display pixel may be subject to process, voltage, and temperature (PVT) variations. Due to such variations, transistor threshold voltages between different display pixels may vary. Variations in transistor threshold voltages can cause the display pixels to produce amounts of light that do not match a desired image. It is within this context that the embodiments herein arise. 
     SUMMARY 
     An electronic device may include a display having an array of display pixels. The display pixels may be organic light-emitting diode display pixels. Each display pixel may include an organic light-emitting diode (OLED) that emits light, a drive transistor coupled in series with the OLED, first and second emission transistors coupled in series with the drive transistor and the OLED, a semiconducting-oxide transistor coupled between gate and drain terminals of the drive transistor, a single storage capacitor coupled to the gate terminal of the drive transistor, a data loading transistor coupled between the source terminal of the drive transistor and a data line, an initialization transistor coupled to the drain terminal of the drive transistor, and an anode reset transistor coupled to the anode terminal of the OLED. The semiconducting-oxide transistor may be an n-type transistor, whereas all remaining transistors in the pixel may be p-type silicon transistors (e.g., PMOS LTPS thin-film transistors). 
     During normal operation, a display pixel may undergo an initialization phase during which the initialization transistor and/or the anode reset transistor is turned on to reset the display pixel. The initialization phase may be followed by one or more on-bias stress phases during which the data loading transistor is activated to load a data voltage at least partially onto the drive transistor. The on-bias stress phase may be automatically followed by a threshold voltage sampling and data loading phase, which is then followed by an emission phase. During the emission phase, the current flowing through the OLED will be independent of the drive transistor threshold voltage due to in-pixel threshold voltage cancellation. 
     Performing the on-bias stress phase prior to the threshold voltage sampling can help mitigate any undesired hysteresis effects and improve first frame response. If desired, the emission phase can be optionally shortened to help reduce potential mismatch between the negative bias temperature stress (NBTS) and the positive bias temperature stress (PBTS) associated with the semiconducting-oxide transistor. If desired, the semiconducting-oxide transistor can also be turned on when the data loading transistor is turned on to lengthen the on-bias stress phase. The display pixel is also operable to support external current sensing (e.g., by turning on the data loading transistor and the initialization transistor) while the display is off or idle. 
     The display pixel may also be configured to support low refresh rate operation (e.g., 1 Hz, 2 Hz, less than 30 Hz, less than 60 Hz, etc.). For low refresh rate operation, a short refresh period is followed by a much longer vertical blanking period. During the refresh period, a first on-bias stress phase may be performed immediately followed by a first threshold voltage sampling and data programming phase; a second on-bias stress phase may be performed after the first threshold voltage sampling and data programming phase; and a third on-bias stress phase may then be performed after the second on-bias stress phase, which is immediately followed by a second threshold voltage sampling and data programming phase. An emission phase can then follow the second threshold voltage sampling and data programming phase. 
     During the vertical blanking period, at least a fourth on-bias stress phase that matches the second on-bias stress phase can be performed to reduce flicker. The initialization voltage may be dynamically adjusted during the second and fourth on-bias stress phases to minimize any potential mismatch. The anode reset voltage may also be dynamically adjusted when switching from the refresh period to the vertical blanking period to help improve low refresh rate performance. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG.  1    is a diagram of an illustrative electronic device having a display in accordance with an embodiment. 
         FIG.  2    is a diagram of an illustrative display having an array of organic light-emitting diode display pixels coupled to compensation circuitry in accordance with an embodiment. 
         FIG.  3    is a circuit diagram of an illustrative display pixel configured to support in-pixel threshold voltage compensation and external threshold voltage compensation in accordance with an embodiment. 
         FIG.  4    is a timing diagram illustrating how multiple on-bias stress operations can be performed with threshold voltage sampling operations to help reduce hysteresis impact in accordance with an embodiment. 
         FIG.  5    is a diagram showing how at least some row control lines can be shared between pixels in adjacent rows in accordance with an embodiment. 
         FIG.  6    is a diagram showing how the on-period for at least some row control signals can be extended to help balance negative bias temperature stress (NBTS) and positive bias temperature stress (PBTS) in accordance with an embodiment. 
         FIG.  7    is a timing diagram illustrating an on-bias stress operation that is optimized to mitigate first frame dimming in accordance with an embodiment. 
         FIG.  8 A  is a diagram showing how a display pixel of the type shown in  FIG.  3    can be configured to support external current sensing operations in accordance with an embodiment. 
         FIG.  8 B  is a timing diagram showing the behavior of relevant row control signals to perform an external current sensing operation in accordance with an embodiment. 
         FIG.  9    is a diagram of a low refresh rate display driving scheme in accordance with an embodiment. 
         FIG.  10    is a circuit diagram of an illustrative display pixel configured to reduce flicker at low refresh rates in accordance with an embodiment. 
         FIG.  11    is a timing diagram illustrating how an initialization voltage can be dynamically adjusted so that the dominant on-bias stress during the refresh phase and the on-bias stress during the vertical blanking phase are well-matched in accordance with an embodiment. 
         FIG.  12    is a timing diagram illustrating how a voltage sampling and data programming operation can be inserted after the dominant on-bias stress to improve first frame response in accordance with an embodiment. 
         FIG.  13    is a timing diagram illustrating how an initialization voltage and/or an anode reset voltage can be dynamically adjusted to match the on-bias stress during the refresh phase and the vertical blanking phase in accordance with an embodiment. 
         FIG.  14 A  is a diagram showing how a display pixel of the type shown in  FIG.  10    can be configured to support external current sensing operations in accordance with an embodiment. 
         FIG.  14 B  is a timing diagram showing the behavior of relevant row control signals to perform an external current sensing operation in accordance with an embodiment. 
         FIGS.  15  and  16    are timing diagrams illustrating other ways of performing on-bias stress during the refresh and vertical blanking phases while dynamically adjusting the initialization and/or anode reset voltage in accordance with certain embodiments. 
         FIG.  17 A  is a circuit diagram of another suitable implementation of an illustrative display pixel circuit in accordance with an embodiment. 
         FIG.  17 B  is a timing diagram illustrating relevant waveforms for operating the pixel circuit shown in  FIG.  17 A  in accordance with an embodiment. 
         FIG.  18 A  is a circuit diagram of yet another suitable implementation of an illustrative display pixel circuit in accordance with an embodiment. 
         FIG.  18 B  is a timing diagram illustrating relevant waveforms for operating the pixel circuit shown in  FIG.  18 A  in accordance with an embodiment. 
         FIG.  19 A  is a circuit diagram of yet another suitable implementation of an illustrative display pixel circuit in accordance with an embodiment. 
         FIG.  19 B  is a timing diagram illustrating relevant waveforms for operating the pixel circuit shown in  FIG.  19 A  in accordance with an embodiment. 
         FIG.  20 A  is a circuit diagram of yet another suitable implementation of an illustrative display pixel circuit in accordance with an embodiment. 
         FIG.  20 B  is a timing diagram illustrating relevant waveforms for operating the pixel circuit shown in  FIG.  20 A  in accordance with an embodiment. 
         FIG.  21 A  is a circuit diagram of yet another suitable implementation of an illustrative display pixel circuit in accordance with an embodiment. 
         FIG.  21 B  is a timing diagram illustrating relevant waveforms for operating the pixel circuit shown in  FIG.  21 A  in accordance with an embodiment. 
     
    
    
     DETAILED DESCRIPTION 
     An illustrative electronic device of the type that may be provided with an organic light-emitting diode (OLED) display is shown in  FIG.  1   . As shown in  FIG.  1   , electronic device  10  may have control circuitry  16 . Control circuitry  16  may include storage and processing circuitry for supporting the operation of device  10 . The storage and processing circuitry may include storage such as hard disk drive storage, nonvolatile memory (e.g., flash memory or other electrically-programmable-read-only memory configured to form a solid-state drive), volatile memory (e.g., static or dynamic random-access-memory), etc. Processing circuitry in control circuitry  16  may be used to control the operation of device  10 . The processing circuitry may be based on one or more microprocessors, microcontrollers, digital signal processors, baseband processors, power management units, audio codec chips, application-specific integrated circuits, programmable integrated circuits, etc. 
     Input-output circuitry in device  10  such as input-output devices  12  may be used to allow data to be supplied to device  10  and to allow data to be provided from device  10  to external devices. Input-output devices  12  may include buttons, joysticks, click wheels, scrolling wheels, touch pads, key pads, keyboards, microphones, speakers, tone generators, vibrators, cameras, sensors, light-emitting diodes and other status indicators, data ports, etc. A user can control the operation of device  10  by supplying commands through input-output devices  12  and may receive status information and other output from device  10  using the output resources of input-output devices  12 . 
     Input-output devices  12  may include one or more displays such as display  14 . Display  14  may be a touch screen display that includes a touch sensor for gathering touch input from a user or display  14  may be insensitive to touch. A touch sensor for display  14  may be based on an array of capacitive touch sensor electrodes, acoustic touch sensor structures, resistive touch components, force-based touch sensor structures, a light-based touch sensor, or other suitable touch sensor arrangements. 
     Control circuitry  16  may be used to run software on device  10  such as operating system code and applications. During operation of device  10 , the software running on control circuitry  16  may display images on display  14  in input-output devices. 
       FIG.  2    shows display  14  and associated display driver circuitry  15 . Display  14  includes structures formed on one or more layers such as substrate  24 . Layers such as substrate  24  may be formed from planar rectangular layers of material such as planar glass layers. Display  14  may have an array of display pixels  22  for displaying images to a user. The array of display pixels  22  may be formed from rows and columns of display pixel structures on substrate  24 . These structures may include thin-film transistors such as polysilicon thin-film transistors, semiconducting oxide thin-film transistors, etc. There may be any suitable number of rows and columns in the array of display pixels  22  (e.g., ten or more, one hundred or more, or one thousand or more). 
     Display driver circuitry such as display driver integrated circuit  15  may be coupled to conductive paths such as metal traces on substrate  24  using solder or conductive adhesive. If desired, display driver integrated circuit  15  may be coupled to substrate  24  over a path such as a flexible printed circuit or other cable. Display driver integrated circuit  15  (sometimes referred to as a timing controller chip) may contain communications circuitry for communicating with system control circuitry  16  over path  125 . Path  125  may be formed from traces on a flexible printed circuit or other cable. Control circuitry  16  (see  FIG.  1   ) may be located on a main logic board in an electronic device such as a cellular telephone, computer, television, set-top box, media player, portable electronic device, or other electronic equipment in which display  14  is being used. 
     During operation, the control circuitry may supply display driver integrated circuit  15  with information on images to be displayed on display  14 . To display the images on display pixels  22 , display driver integrated circuit  15  may supply clock signals and other control signals to display driver circuitry such as row driver circuitry  18  and column driver circuitry  20 . For example, data circuitry  13  may receive image data and process the image data to provide pixel data signals to display  14 . The pixel data signals may be demultiplexed by column driver circuitry  20  and pixel data signals D may be routed to each pixel  22  over data lines  26  (e.g., to each red, green, or blue pixel). Row driver circuitry  18  and/or column driver circuitry  20  may be formed from one or more integrated circuits and/or one or more thin-film transistor circuits. 
     Display driver integrated circuit  15  may include compensation circuitry  17  that helps to compensate for variations among display pixels  22  such as threshold voltage variations. Compensation circuitry  17  may, if desired, also help compensate for transistor aging. Compensation circuitry  17  may be coupled to pixels  22  via path  19 , switching circuitry  21 , and paths  23 . Compensation circuitry  17  may include sense circuitry  25  and bias circuitry  27 . Sense circuitry  25  may be used in sensing (e.g., sampling) voltages from pixels  22 . During sense operations, switching circuitry  21  may be configured to electrically couple sense circuitry  25  to one or more selected pixels  22 . For example, compensation circuitry  17  may produce control signal CTL to configure switching circuitry  21 . Sense circuitry  25  may sample currents, voltages or other desired signals from the pixels over path  19 , switching circuitry  21 , and paths  23 . Bias circuitry  27  may include one or more driver circuits for driving reference or bias voltages onto nodes of pixels  22 . For example, switching circuitry  21  may be configured to electrically couple path  19  to one or more selected pixels  22 . In this scenario, bias circuitry  27  may provide reference signals to the selected pixels. The reference signals may bias nodes at the selected pixels at desired voltages for the sensing operations performed by sense circuitry  25 . 
     Compensation circuitry  17  may perform compensation operations on pixels  22  using bias circuitry  27  and sense circuitry  25  to generate compensation data that is stored in storage  29 . Storage  29  may, for example, be static random-access memory (SRAM). In the example of  FIG.  2   , storage  29  is on-chip storage. If desired, storage  29  may be off-chip storage such as non-volatile storage (e.g., non-volatile memory that maintains stored information even when the display is powered off). The compensation data stored in storage  29  may be retrieved by data circuitry  13  during display operations. Data circuitry  13  may process the compensation data along with incoming digital image data to generate compensated data signals for pixels  22 . 
     Data circuitry  13  may include gamma circuitry  44  that provides a mapping of digital image data to analog data signals at appropriate voltage levels for driving pixels  22 . Multiplexer  46  receives a set of possible analog data signals from gamma circuitry  44  and is controlled by the digital image data to select an appropriate analog data signal for the digital image data. Compensation data retrieved from storage  29  may be added to (or subtracted from) the digital image data by adder circuit  48  to help compensate for transistor variations (e.g., threshold voltage variations, transistor aging variations, or other types of variations) between different display pixels  22 . This example in which compensation data is added as an offset to digital input image data is merely illustrative. In general, data circuitry  13  may process compensation data along with image data to produce compensated analog data signals for driving pixels  22 . 
     In contrast to techniques that focus on performing in-pixel threshold canceling (such as by performing an initialization phase followed by a threshold sampling phase), performing sensing and compensation in this way using compensation circuitry  17  outside of each pixel  22  allows for higher refresh rates (e.g., greater than 60 Hz refresh rate, at least 120 Hz refresh rate, etc.) and is sometimes referred to as “external” compensation. External variation compensation may be performed in the factory, in real time (e.g., during blanking intervals between successive image frames), or when the display is idle (as examples). In accordance with at least some embodiments, display  14  may be operated using a hybrid compensation scheme in which in-pixel threshold canceling is implemented during normal display operation and external threshold compensation is implemented while display  14  is turned off Configured in this way, the in-pixel compensation can help mitigate threshold voltage hysteresis (which improves first frame response), whereas the external compensation can help mitigate aging and other transistor reliability issues. 
     Row driver circuitry  18  may be located on the left and right edges of display  14 , on only a single edge of display  14 , or elsewhere in display  14 . During operation, row driver circuitry  18  may provide row control signals on horizontal lines  28  (sometimes referred to as row lines, “scan” lines, and/or “emission” lines). Row driver circuitry  18  may include scan line driver circuitry for driving the scan lines and emission line driver circuitry for driving the emission lines. The scan line and emission line driver circuitry may sometimes be referred to as gate driver circuitry. 
     Demultiplexing circuitry  20  may be used to provide data signals D from display driver integrated circuit (DIC)  15  onto a plurality of corresponding vertical lines  26 . Demultiplexing circuitry  20  may sometimes be referred to as column driver circuitry, data line driver circuitry, or source driver circuitry. Vertical lines  26  are sometimes referred to as data lines. During display operations, display data may be loaded into display pixels  22  using lines  26 . 
     Each data line  26  is associated with a respective column of display pixels  22 . Sets of horizontal signal lines  28  run horizontally across display  14 . Each set of horizontal signal lines  28  is associated with a respective row of display pixels  22 . The number of horizontal signal lines in each row is determined by the number of transistors in the display pixels  22  that are being controlled independently by the horizontal signal lines. Display pixels of different configurations may be operated by different numbers of scan lines. 
     Row driver circuitry  18  may assert control signals such as scan and emission signals on the row lines  28  in display  14 . For example, driver circuitry  18  may receive clock signals and other control signals from display driver integrated circuit  15  and may, in response to the received signals, assert scan control signals and an emission control signal in each row of display pixels  22 . Rows of display pixels  22  may be processed in sequence, with processing for each frame of image data starting at the top of the array of display pixels and ending at the bottom of the array (as an example). While the scan lines in a row are being asserted, control signals and data signals that are provided to column driver circuitry  20  by DIC  15  may direct column driver circuitry  20  to demultiplex and drive associated data signals D (e.g., compensated data signals provided by data circuitry  13 ) onto data lines  26  so that the display pixels in the row will be programmed with the display data appearing on the data lines D. The display pixels can then display the loaded display data. 
     The external pixel compensation scheme described above may involve using sense circuitry  25  to perform current sensing on selected display pixels. In general, the amount of emission current flowing through each display pixel is dependent on the threshold voltage of a “drive” thin-film transistor (TFT) within that display pixel. The threshold voltage of the drive transistor may also vary depending on the current value of the gate-to-source voltage Vgs of the drive transistor. For example, the drive transistor threshold voltage may exhibit a first average level when Vgs is being raised from low to high, but may exhibit a second average level that is different than the first average level when Vgs is being lowered from high to low, thus yielding different current-voltage (I-V) characteristic curves. This dependence of the threshold voltage on the actual Vgs value is sometimes referred to as transistor “hysteresis,” and if care is not taken, this hysteresis can negatively impact the accuracy of the current sensing operations performing by circuitry  25 . 
       FIG.  3    is a circuit diagram of an illustrative organic light-emitting diode display pixel  22  in display  14  that is operable to support both in-pixel threshold voltage compensation and external threshold voltage compensation. As shown in  FIG.  3   , display pixel  22  may include a storage capacitor Cst, an n-type (i.e., n-channel) transistor such as semiconducting-oxide transistor Toxide, and p-type (i.e., p-channel) transistors such as a drive transistor Tdrive, a data loading transistor Tdata, a first emission transistor Tem 1 , second emission transistor Tem 2 , a first initialization transistor Tini 1 , and a second initialization transistor Tini 2 . While transistor Toxide is formed using semiconducting oxide (e.g., a transistor with a channel formed from semiconducting oxide such as indium gallium zinc oxide or IGZO), the other p-channel transistors may be thin-film transistors formed from a semiconductor such as silicon (e.g., polysilicon channel deposited using a low temperature process, sometimes referred to as LTPS or low-temperature polysilicon). Semiconducting-oxide transistors exhibit relatively lower leakage than silicon transistors, so implementing transistor Toxide as a semiconducting-oxide transistor will help reduce flicker (e.g., by preventing current from leaking away from the gate terminal of drive transistor Tdrive). 
     In another suitable arrangement, transistors Toxide and Tdrive may be implemented as semiconducting-oxide transistors while the remaining transistors Tdata, Tem 1 , Tem 2 , Tini 1 , and Tini 2  are LTPS transistors. Transistor Tdrive serves as the drive transistor and has a threshold voltage that is critical to the emission current of pixel  22 . Since the threshold voltage of transistor Tdrive may experience hysteresis, forming the drive transistor as a top-gate semiconducting-oxide transistor can help reduce the hysteresis (e.g., a top-gate IGZO transistor experiences less Vth hysteresis than a silicon transistor). If desired, any of the remaining transistors Tdata, Tem 1 , Tem 2 , Tini 1 , and Tini 2  may be implemented as semiconducting-oxide transistors. Moreover, any one or more of the p-channel transistors may be n-type (i.e., n-channel) thin-film transistors. 
     Display pixel  22  may include an organic light-emitting diode (OLED)  304 . A positive power supply voltage VDDEL may be supplied to positive power supply terminal  300 , and a ground power supply voltage VSSEL may be supplied to ground power supply terminal  302 . Positive power supply voltage VDDEL may be 3 V, 4 V, 5 V, 6 V, 7 V, 2 to 8 V, or any suitable positive power supply voltage level. Ground power supply voltage VSSEL may be 0 V, −1 V, −2 V, −3 V, −4 V, −5 V, −6V, −7 V, or any suitable ground or negative power supply voltage level. The state of drive transistor Tdrive controls the amount of current flowing from terminal  300  to terminal  302  through diode  304 , and therefore the amount of emitted light  306  from display pixel  22 . Organic light-emitting diode  304  may have an associated parasitic capacitance CO LED  (not shown). 
     Terminal  308  may be used to supply an initialization voltage Vini (e.g., a negative voltage such as −1 V, −2 V, −3 V, −4V, −5 V, −6 V, or other suitable voltage) to assist in turning off diode  304  when diode  304  is not in use. Terminal  308  is therefore sometimes referred to as the initialization line. Control signals from display driver circuitry such as row driver circuitry  18  of  FIG.  2    are supplied to control terminals such as row control terminals  312 ,  314 - 1 ,  314 - 2 , and  314 - 2 ′. Row control terminal  312  may serve as an emission control terminal (sometimes referred to as an emission line or emission control line), whereas row control terminals  314 - 1  and  314 - 2  may serve as first and second scan control terminals (sometimes referred to as scan lines or scan control lines). Emission control signal EM may be supplied to terminal  312 . Scan control signals Scan 1  and Scan 2  may be applied to scan terminals  314 - 1  and  314 - 2 , respectively. Scan control signal Scan 2  from a preceding row in the array of display pixels may be applied to scan terminal  314 - 2 ′. A data input terminal such as data signal terminal  310  is coupled to a respective data line  26  of  FIG.  1    for receiving image data for display pixel  22 . Data terminal  310  may also be referred to as the data line. 
     Control signals EM(n), Scan 2 ( n ), and Scan 2 ( n −1) for modulating the p-type silicon transistors can be driven low to turn on those transistors (since p-type transistors are “active-low” devices) and driven high to turn them on. Control signals EM(n), Scan 2 ( n ), and Scan 2 ( n −1), when asserted, may generally be driven to a voltage level that is lower than VSSEL (e.g., to overdrive the corresponding transistors). As an example, if VSSEL is equal to −3.5 V, signals EM(n), Scan 2 ( n ), and Scan 2 ( n −1) might be driven to −9 V when asserted. Control signals EM(n), Scan 2 ( n ), and Scan 2 ( n −1), when deasserted, may generally be driven to a voltage level that is higher than VDDEL (e.g., to further deactivate the corresponding transistors to help minimize leakage). As an example, if VDDEL is equal to 4.5 V, signals EM(n), Scan 2 ( n ), and Scan 2 ( n −1) might be driven to 7 V when deasserted. 
     Control signal Scan 1 ( n ) for modulating the n-type semiconducting-oxide transistor Toxide can be driven high to turn on transistor Toxide (since n-type transistors are “active-high” devices) and driven low to turn off transistor Toxide. Since Scan 1  independently controls transistor Toxide, the high and low levels of Scan 1  can be adjusted to enhance oxide TFT driving capability. Control signal Scan 1 ( n ), when asserted, may generally be driven to a voltage level that is higher than VDDEL to overdrive transistor Toxide. As an example, if VDDEL is equal to 5 V, signal Scan 1 ( n ) might be driven to 12 V when asserted. Control signal Scan 1 ( n ), when deasserted, may generally be driven to a voltage level that is lower than VSSEL to minimize leakage through transistor Toxide. As an example, if VSSEL is equal to −2 V, signal Scan 1 ( n ) might be driven to −6 V when deasserted. The disclosed high and low voltage levels for each of these row control signals are merely illustrative and can be adjusted to other suitable voltage levels to support the desired mode of operation. 
     In the example of  FIG.  3   , transistors Tem 1 , Tdrive, Tem 2 , and OLED  304  may be coupled in series between power supply terminals  300  and  302 . In particular, first emission control transistor Tem 1  may have a source terminal that is coupled to positive power supply terminal  300 , a gate terminal that receives emission control signal EM(n) via emission line  312 , and a drain terminal (labeled as Node 1 ). The notation “(n)” indicates that the corresponding signal is generated using a gate driver associated with that row of display pixels. The terms “source” and “drain” terminals of a transistor can sometimes be used interchangeably and may therefore sometimes be referred to as “source-drain” terminals. 
     Drive transistor Tdrive may have a source terminal coupled to Node 1 , a gate terminal (labeled as Node 2 ), and a drain terminal (labeled as Node 3 ). Second emission control transistor Tem 2  may have a source terminal coupled to Node 3 , a gate terminal that also receives emission control signal EM(n) via emission line  312 , and a drain terminal (labeled as Node 4 ) coupled to ground power supply terminal  302  via light-emitting diode  304 . Configured in this way, emission control signal EM(n) can be asserted (e.g., driven low or temporarily pulsed low) to turn on transistors Tem 1  and Tem 2  during an emission phase to allow current to flow through light-emitting diode  304 . 
     Storage capacitor Cst may have a first terminal that is coupled to positive power supply line  300  and a second terminal that is coupled to Node 2 . Image data that is loaded into pixel  22  can be at least be partially stored on pixel  22  by using capacitor Cst to hold charge throughout the emission phase. Transistor Toxide may have a source terminal coupled to Node 2 , a gate terminal configured to receive scan control signal Scan 1 ( n ) via scan line  314 - 2 , and a drain terminal coupled to Node 3 . Signal Scan 1 ( n ) may be asserted (e.g., driven high or temporarily pulsed high) to turn on n-type transistor Toxide to short the drain and gate terminals of transistor Tdrive. A transistor configuration where the gate and drain terminals are shorted is sometimes referred to as being “diode-connected.” 
     Data loading transistor Tdata may have a source terminal coupled to data line  310 , a gate terminal configured to receive scan control signal Scan 2 ( n ) via scan line  314 - 2 , and a drain terminal coupled to Node 1 . Configured in this way, signal Scan 2 ( n ) can be asserted (e.g., driven low or temporarily pulsed low) to turn on transistor Tdata, which will allow a data voltage from data line  310  to be loaded onto Node 1 . 
     Transistor Tini 1  may have a source terminal coupled to Node 3 , a gate terminal configured to receive scan control signal Scan 2 ( n −1) via scan line  314 - 2 ′, and a drain terminal coupled to initialization line  308 . The notation “(n−1)” indicates that the corresponding signal is generated using a gate driver associated with a preceding row of display pixels (e.g., Scan 2 ( n −1) represents the Scan 2  signal that controls transistors Tdata in the immediately preceding row). Transistor Tini 2  may have a source terminal coupled to Node 4 , a gate terminal configured to receive scan control signal Scan 2 ( n −1) via scan line  314 - 2 ′, and a drain terminal coupled to initialization line  308 . Configured in this way, scan control signal Scan 2 ( n −1) can be asserted (e.g., driven low or temporarily pulsed low) to turn on transistors Tini 1  and Tini 2 , which drives both Node 3  and Node 4  down to initialization voltage Vini. 
     During normal data refresh period, display pixel  22  may be operated in at least four different types of phases: (1) an initialization/reset phase, (2) an on-bias stress phase, (3) a threshold voltage sampling and data writing phase, and (4) an emission phase—not necessarily in this order.  FIG.  4    is a timing diagram showing relevant signal waveforms that may be applied to display pixel  22  during normal operation. 
     Prior to time t 1 , only signal EM(n) is asserted so pixel  22  is in the emission phase. At time t 1 , signal EM(n) is deasserted or driven low, which marks the end of the emission phase. At time t 2  (at the beginning of the initialization phase), control signals Scan 1 ( n ) and Scan 2 ( n −1) are asserted. Asserting signal Scan 2 ( n −1) will turn on transistors Tini 1  and Tini 2  in parallel, which will drive Node 3  and Node 4  to Vini. Node 3  is at the drain terminal of transistor Tdrive, so the corresponding voltage Vd at Node 3  will be initialized to Vini during this time (i.e., Vd=Vini). Since Node 4  is at the anode terminal of light-emitting diode  304 , setting Node 4  to Vini is sometimes referred to as performing “anode reset.” Asserting signal Scan 1 ( n ) will turn on transistor Toxide, which shorts the gate and drain terminals of transistor Tdrive and therefore pulls the voltage at the gate terminal of the drive transistor Vg also down to Vini. During the initialization phase, the voltage across capacitor Cst is therefore reset to a predetermined voltage difference (VDDEL−Vini). 
     Signal Scan 2 ( n −1) is deasserted at time t 3  to turn off transistors Tini 1  and Tini 2 , which marks the end of the initialization and anode reset phase. Signal Scan 1 ( n ) may remain asserted until the subsequent emission phase (e.g., transistor Toxide will remain on during the entirety of the initialization phase and the threshold voltage sampling and data writing phases). 
     At time t 4 , signal Scan 2 ( n ) is pulsed low to temporarily activate data loading transistor Tdata. Turning on transistor Tdata will load a data voltage Vdata onto the source terminal of the drive transistor such that the voltage Vs at Node 1  is set to Vdata (i.e., Vs=Vdata). Since the drive transistor is currently in the diode-connected configuration (because Toxide is turned on), the drive transistor will pull gate voltage Vg up to (Vdata−Vth), where Vth represents the threshold voltage of the drive transistor. Thus, the voltage across capacitor Cst is now set to (VDDEL−Vdata+Vth). As such, drive transistor threshold voltage Vth has been successfully sampled and Vdata has been successfully programmed/written onto storage capacitor Cst. 
     The assertion of signal Scan 2 ( n ) at time t 4  sets Vs to Vdata, which will then prompt the drive transistor to pull its gate voltage Vg from Vini up towards (Vdata−Vth). This brief period of time (see shaded portion in  FIG.  4   ) while Vg is charging up to (Vdata−Vth) represents an on-bias stress phase. At the beginning of the on-bias stress phase (i.e., at time t 4 ), the source-to-gate voltage of the drive transistor Vsg may be equal to (Vdata−Vini) so that Vdata is at least partially applied to the drive transistor prior to any threshold voltage sampling. Applying Vdata to pixel  22  prior to any threshold voltage sampling may be technically advantageous for the following reasons. 
     In certain situations, threshold voltage Vth can shift, such as when display  14  is transitioning from a black image to a white image or when transitioning from one gray level to another. This shifting in Vth (sometimes referred to herein as thin-film transistor “hysteresis”) can cause a reduction in luminance, which is otherwise known as “first frame dimming.” For example, the saturation current Ids waveform as a function of Vgs of the drive transistor for a black frame might be slightly offset from the target Ids waveform as a function of Vgs of the drive transistor for a white frame. Without performing on-bias stress, the sampled Vth will correspond to the black frame and will therefore deviate from the target Ids waveform by quite a large margin. By performing on-bias stress, the sampled Vth will correspond to Vdata and will therefore be much closer to the target Ids curve. Performing the on-bias stress phase to bias the Vsg of the drive transistor with Vdata before sampling Vth can therefore help mitigate hysteresis and improve first frame response. An on-bias stress phase may therefore be defined as an operation that applies a suitable bias voltage directly to the drive transistor during non-emission phases (e.g., such as by turning on the data loading transistor or the initialization transistor). Thus, although  FIG.  4    shows Vth sampling and data writing phase as beginning at time t 4 , only the OBS phase begins at time t 4 , and the Vth sampling and data programming occurs immediately after the OBS phase (e.g., OBS will be followed automatically by the Vth sampling and data writing operation without having turning on any other transistors in pixel  22 ). 
     At time t 5 , signal Scan 2 ( n ) is deasserted, which marks the end of the Vth sampling the data programming phase. As shown in  FIG.  4   , the on-bias stress phase has a relatively short duration compared to the rest of the Vth sampling and data programming phase. To ensure the efficacy of the on-bias stress, signal Scan 2 ( n ) can be pulsed multiple times to perform additional on-bias stress operations. In the example of  FIG.  4   , signal Scan 2 ( n ) is pulsed low from time t 6  to t 7  to trigger a second on-bias stress phase and a second Vth sampling and data programming phase and is again pulsed low from time t 8  to t 9  to trigger a third on-bias stress phase and a third Vth sampling and data programming phase. The data loaded during the final data programming phase (see, e.g., data signal D(n)) represents the actual data value that will be displayed by this display pixel. The example of  FIG.  4    in which three separate on-bias stress phases are performed is merely illustrative. If desired, less than three or more than three on-bias stress phases may be provided to help reduce the impact of Vth hysteresis. 
     At time t 10 , emission control signal EM(n) may again be asserted to signify the beginning of the emission phase. Asserting signal EM(n) will turn on transistors Tem 1  and Tem 2 , which will pull Vs up to VDDEL. The resulting source-to-gate voltage Vsg of transistor Tdrive will be equal to VDDEL−(Vdata−Vth). Since the final emission current is proportional to Vsg minus Vth, the emission current will be independent of Vth since (Vsg−Vth) will be equal to (VDDEL−Vdata+Vth−Vth), where Vth cancels out. This type of operating scheme where the drive transistor threshold voltage is internally sampled and canceled out in this way is sometimes referred to as in-pixel threshold voltage compensation. 
     In general, each of the row control signals is associated with only one of the rows in the array of display pixels. In certain embodiments, some of the row control lines can be shared between display pixels in adjacent rows (see, e.g.,  FIG.  5   ). As shown in  FIG.  5   , gate driver circuitry such gate driver stage  500  may drive row control signals EM and Scan 1  that are shared between pixels in two neighboring rows and may also drive signal Scan 2 (2n−1) that is fed only to the first (odd) row of pixels  22  and signal Scan 2 (2n) that is fed only to second (even) row of pixels  22 . Gate driver stage  500  may represent one stage in a chain of stages in row driver circuitry  18  (see  FIG.  2   ). While signals Scan 1  and EM can be shared among multiple adjacent rows, signal Scan 2  cannot be shared since it controls the data loading (e.g., different pixels need to be loaded with different data signals to maintain full display resolution). 
     In the exemplary operation of  FIG.  4   , the duration for which signal Scan 1  is high may be much shorter than the duration for which signal Scan 1  is low (i.e., the emission phase is much longer than the non-emission phases). Signal Scan 1  directly controls transistor Toxide within pixel  22 . When signal Scan 1  is low, transistor Toxide is turned off and is subject to negative bias temperature stress (NBTS). When signal Scan 1  is high, transistor Toxide is turned on and is subject to positive bias temperature stress (PBTS). NBTS can cause oxide transistor threshold voltage Vth to shift in the negative direction over time, whereas PBTS can cause Vth to shift in the positive direction over time. When the emission phases are much longer than the non-emission phases, NBTS will dominate and may cause a negative drift in Vth over the lifetime of transistor Toxide, which also degrades the reliability of that transistor. 
     To help improve the reliability of the oxide transistor, the duration for which signal Scan 1  is high may be adjusted, lengthened, or optimized to help balance the NBTS and PBTS (see, e.g.,  FIG.  6   ). In the timing diagram of  FIG.  6   , the period during which signal Scan 1  is asserted may be extended as shown by dotted portion  600 . When signal Scan 1 ( n ) is asserted, signals Scan 2 ( n −1) and Scan 2 ( n ) may be pulses at least two times (as shown in  FIG.  6   ), more than two times, three or more times, four to ten times, 10 or more times, 100 or more times, or any suitable number of times to perform the on-bias stress and Vth sampling and data programming operations. By tuning the on-period of the oxide transistor relative to its off-period, the risk of Vth shift can be minimized and the oxide TFT lifetime can be improved. 
       FIG.  7    is a timing diagram illustrating how the on-bias stress phase can be further optimized to mitigate first frame dimming in accordance with another suitable arrangement. In contrast to the example of  FIG.  4    in which signal Scan 1  is constantly asserted during the non-emission period,  FIG.  7    illustrates how signal Scan 1  can be pulsed low during the non-emission period to provide an enhanced on-bias stress effect. 
     Prior to time t 1 , only signal EM(n) is asserted, so pixel  22  is in the emission phase. At time t 1 , signal EM(n) is deasserted or driven low, which marks the end of the emission phase. Signal Scan 1 ( n ) is asserted some time after t 1 , which turns on transistor Toxide. At time t 2  (at the beginning of the initialization phase), control signal Scan 2 ( n −1) is asserted or pulsed low. Asserting signal Scan 2 ( n −1) will turn on transistors Tini 1  and Tini 2  in parallel, which will drive Node 3  and Node 4  to Vini. Node 3  is at the drain terminal of transistor Tdrive, so the corresponding voltage Vd at Node 3  will be initialized to Vini during this time (i.e., Vd=Vini). The OLED anode terminal Node 4  will also be reset to Vini. Since signal Scan 1 ( n ) is asserted, transistor Toxide will be on, which shorts the gate and drain terminals of transistor Tdrive and therefore pulls the voltage at the gate terminal of the drive transistor Vg also down to Vini. During the initialization and anode reset phase, the voltage across capacitor Cst is therefore reset to a predetermined voltage difference (VDDEL−Vini). 
     Signal Scan 2 ( n −1) is deasserted at time t 3  to turn off transistors Tini 1  and Tini 2 , which marks the end of the initialization and anode reset phase. Signal Scan 1 ( n ) may remain asserted until the subsequent emission phase (e.g., transistor Toxide will remain on during the entirety of the initialization phase and the threshold voltage sampling and data writing phases). 
     At time t 4 , signal Scan 1 ( n ) is deasserted or pulsed low. Driving signal Scan 1 ( n ) low will turn off transistor Toxide so that the gate and drain terminals of the drive transistor Tdrive is no longer shorted (i.e., so that the drive transistor is no longer diode-connected). At time t 4 , control signal Scan 2 ( n ) is also pulsed low, which turns on data loading transistor Tdata and sets the source terminal voltage Vs to Vdata. Since the oxide transistor is turned off, gate terminal voltage Vg stays at initialization voltage Vini, which will cause the drain terminal voltage Vd to be pulled up to Vdata. Note that while transistor Toxide is turned off, no in-pixel Vth sampling can occur, so the entire duration from time t 4  to t 5  will serve as the on-bias stress phase. This period during which Scan 1 ( n ) is pulsed low from time t 4  to t 5  can be adjusted or optimized to improve first frame response of the display. Extending the on-bias stress phase in this way can also help obviate the need to perform multiple smaller on-bias stress operations as shown in the example of  FIG.  4   , which can reduce dynamic power consumption. At time t 5 , scan signal Scan 1 ( n ) is reasserted, which turns on transistor Toxide. 
     At time t 6 , control signal Scan 2 ( n ) is asserted or pulsed low, which sets Vs to Vdata. Since the drive transistor is currently in the diode-connected configuration (because Toxide is enabled), the drive transistor will pull gate voltage Vg up to (Vdata−Vth). Thus, the voltage across capacitor Cst is now set to (VDDEL−Vdata+Vth). As such, drive transistor threshold voltage Vth has been successfully sampled and Vdata has been successfully programmed/written onto storage capacitor Cst. At time t 7 , signal Scan 2 ( n ) is deasserted, which marks the end of the Vth sampling and data programming phase. 
     At time t 8 , emission control signal EM(n) may again be asserted to signify the beginning of the emission phase. Asserting signal EM(n) will turn on transistors Tem 1  and Tem 2 , which will pull Vs up to VDDEL. The resulting source-to-gate voltage Vsg of transistor Tdrive will be equal to VDDEL−(Vdata−Vth). Since the final emission current is proportional to Vsg minus Vth, the emission current will be independent of Vth since (Vsg−Vth) will be equal to (VDDEL−Vdata+Vth−Vth), where Vth cancels out to achieve the in-pixel threshold voltage compensation. 
     In addition to performing the “in-pixel” threshold canceling described above in connection with  FIG.  4    or  FIG.  7   , “external” threshold voltage compensation may also be performed using compensation circuitry  17  outside of each pixel  22 . External variation compensation may, for example, be performed in the factory, when the display is idle or turned off, or in real time (e.g., during blanking intervals between successive image frames). While in-pixel threshold voltage compensation helps reduce hysteresis, external threshold voltage compensation can help mitigate transistor aging, drive transistor Vth shift over the lifetime of the display pixel, and other TFT reliability issues. An operating scheme where both in-pixel and external Vth compensation are achieved is sometimes referred to as a “hybrid” threshold voltage compensation driving scheme. 
       FIG.  8 A  is a diagram showing how a display pixel of the type shown in  FIG.  3    can be configured to support external current sensing operations.  FIG.  8 B  is a timing diagram showing the behavior of relevant row control signals to perform such external current sensing operations. As shown in  FIG.  8 B , the odd row scan control signal Scan 2 _odd(n) can be pulsed low to perform the initialization and anode reset phase, and the even row scan control signal Scan 2 _even(n) can then be pulsed low to perform the Vth sampling and data programming phase. At some later time (e.g., when the display is turned off/idle or during some other time when the user is not viewing the display), both the even and odd Scan 2  control signals are simultaneously asserted while Scan 1  is deasserted to perform the current sensing operations. 
     Referring back to  FIG.  8 A , a low Scan 1  ( n ) turns off transistor Toxide, whereas a low Scan 2 _even(n) and a low Scan 2 _odd(n) will turn on transistors Tdata and Tini 1 . Emission control signal EM(n) should be deasserted during this time, which deactivates transistors Tem 1  and Tem 2 . Configured in this way, a sensing current can flow from data line  310  through transistors Tdata, Tdrive, and Tini 1  onto initialization line  308 , as indicated by sensing current path  800 . Current  800  can be measured using sense circuitry  25  (see  FIG.  2   ) to generate compensation data that is stored in storage circuitry  29 . As described above, external Vth compensation via current sensing in combination with in-pixel Vth canceling can help minimize any undesired TFT effects associated with the threshold voltage of the drive transistor, which helps maintain consistent luminance levels over the lifespan of the display. 
     Display  14  may optionally be configured to support low refresh rate operation. Operating display  14  using a relatively low refresh rate (e.g., a refresh rate of 1 Hz, 2 Hz, 1-10 Hz, less than 30 Hz, less than 60 Hz, or other low rate) may be suitable for applications outputting content that is static or nearly static and/or for applications that require minimal power consumption.  FIG.  9    is a diagram of a low refresh rate display driving scheme in accordance with an embodiment. As shown in  FIG.  9   , display  14  may alternate between a short data refresh period (as indicated by period T_refresh) and an extended vertical blanking period (as indicated by period T_blank). As an example, each data refresh period T_refresh may be approximately 16.67 milliseconds (ms) in accordance with a 60 Hz data refresh operation, whereas each vertical blanking period T_blank may be approximately 1 second so that the overall refresh rate of display  14  is lowered to 1 Hz. Configured as such, refresh duration T_blank can be adjusted to tune the overall refresh rate of display  14 . For example, if the duration of T_blank was tuned to half a second, the overall refresh rate would be increased to approximately 2 Hz. In the embodiments described herein, T_blank may be at least two times, at least ten times, at least 30 times, or at least 60 times longer in duration than T_refresh (as examples). 
     A schematic diagram of an illustrative organic light-emitting diode display pixel  22  in display  14  that can be used to support low refresh rate operation is shown in  FIG.  10   . Pixel  22  of  FIG.  10    may have similar structure as pixel  22  shown in  FIG.  3    (i.e., pixel  22  in  FIG.  10    has the same number of transistors and capacitor as pixel  22  in  FIG.  3   ). Emission transistors Tem 1  and Tem 2  have gates configured to receive emission control signal EM(n) via emission line  312 . Semiconducting-oxide transistor Toxide has a gate terminal configured to receive a first scan control signal SC 1 ( n ) via first scan line  314 - 1 . Data loading transistor Tdata has a gate terminal configured to receive second scan control signal SC 2 ( n ) via second scan line  314 - 2 . 
     In contrast to the pixel configuration of  FIG.  3   , the initialization line in display pixel  22  of  FIG.  10    is only connected to one transistor within pixel  22 . As shown in  FIG.  10   , an initialization transistor Tini has a source terminal coupled to Node 3  (i.e., the drain terminal of the drive transistor), a gate terminal configured to receive a third scan control signal SC 3 ( n ) via third scan line  314 - 3 , and a drain terminal coupled to a dynamic initialization line  308 ′. Display pixel  22  may further include an anode reset transistor Tar that has a source terminal coupled to Node 4  (i.e., the anode terminal of OLED  304 ), a gate terminal configured to receive scan control signal SC 3 ( n +1) generated from a subsequent row in the array, and a drain terminal coupled to an anode reset line  309 . Dynamic initialization line  308 ′ and anode reset line  309  may be separate control lines such that the initialization voltage Vdini(n) on line  308 ′ and the anode reset voltage Var on line  309  can be biased to different levels during operation of pixel  22 . 
       FIG.  11    is a timing diagram showing relevant signal waveforms that may be applied to display pixel  22  of  FIG.  10    during both the refresh period (sometimes referred to as the “refresh frame”) and the vertical blanking period (sometimes referred to as the “vertical blanking frame”). From time t 1  to t 2 , scan control signals SC 1 ( n ) and SC 3 ( n ) are asserted to perform the initialization phase. During the initialization phase, the initialization line  308 ′ is biased at low voltage VL, which will cause drain voltage Vd at Node 3  will be pulled down to voltage VL. Since transistor Toxide is also on during this time, gate voltage Vg at Node 2  will also be pulled down to VL. As a result, the voltage across capacitor Cst will be set to a predetermined voltage difference (VDDEL−VL). 
     At time t 3 , scan control signal SC 2 ( n ) will be pulsed low to perform the Vth sampling and data writing phase. As described above in connection with  FIG.  3   , the brief period of time following the assertion of signal SC 2 ( n ) where the drive transistor is temporarily activated to charge gate voltage Vg up to (Vdata−Vth) represents a first on-bias stress phase OBS 1 . By the end of time t 4 , the source voltage Vs at Node 1  will be set to Vdata, whereas Vg and Vd will both be set to (Vdata−Vth) by virtue of the diode connection of the drive transistor. Thus, the voltage across storage capacitor Cst will be set of (VDDEL−Vdata+Vth). 
     In low refresh rate operation, the vertical blanking frame may be much longer than the refresh frame. To prevent Vth drift during the vertical blanking frame, it would be desirable to also implement one or more on-bias stress phases during the vertical blanking frame. During the vertical blanking frame, however, signals SC 1 ( n ) and SC 2 ( n ) cannot be asserted to turn on charge up Vs and Vd as a function of Vdata. Thus, another mechanism must be introduced to charge up Vs and Vd. In accordance with an embodiment, initialization voltage Vdini(n) may be dynamically raised from low voltage VL to a high voltage VH while asserting signal SC 3 ( n ) to perform a pseudo on-bias stress phase OBS 2 ′ during the vertical blanking frame. Voltage VH may be at least equal to or greater than Vdata, which will turn on drive transistor (whose gate is held at (Vdata−Vth) by capacitor Cst) and ensure that voltage Vs at Node 1  is also charged to VH. 
     Initialization voltage Vdini may be dynamically adjusted on a per-row basis, so signal Vdini(n) is a row-based signal (e.g., signal Vdini may be asserted at different times for different rows). In contrast, the anode reset voltage Var may be a fixed direct current (DC) global voltage signal. The example of  FIG.  11    in which one on-bias stress operation OBS 2 ′ is performed during the vertical blanking frame is merely illustrative. In general, two or more on-bias stress operations OBS 2 ′ may be performed during the vertical blanking frame. For example, on-bias stress operations OBS 2 ′ may be performed at a relatively high frequency of 30 Hz, 60 Hz, 120 Hz, 240 Hz, 10-240 Hz, or other suitable frequency. 
     By inspection, on-bias stress phase OBS 2 ′ from time t 7  to t 8  is qualitatively different than on-bias stress phase OSB 1  from time t 3  to t 4  (i.e., the duration of the on-bias stress will be different, and the actual voltage applied to the source-drain terminals of the drive transistor will also be different). This mismatch in OBS 1  versus OBS 2 ′ might create noticeable flicker. 
     To help reduce flicker, an additional on-bias stress phase OBS 2  can be inserted between the Vth sampling and data programming phase and the emission phase (see, e.g., OBS 2  inserted from time t 5  to t 6 ). As shown in  FIG.  11   , additional on-bias stress phase OBS 2  in the refresh frame may be qualitatively identical to that of OBS 2 ′ in the vertical blanking frame. For example, signal SC 3 ( n ) may be pulsed low for the same duration, signal Vdini(n) may be dynamically biased to the same VH level, and the duration t 5 -t 6  may be equal to the duration t 7 -t 8 ). The longer on-bias stress phase OBS 2  will dominate the prior/shorter on-bias stress phase OBS 1 , and reducing on-bias stress mismatch between the refresh and vertical blanking periods will provide improved display flicker performance. 
     In the example of  FIG.  11   , on-bias stress phase OBS 2  is inserted immediately prior to the emission phase. In certain scenarios, it is possible for drive transistor threshold Vth to shift during the inserted phase OBS 2 . For instance, hysteresis and temperature variation could cause Vth to shift during OBS 2 , which can lead to undesirable Mura effects and degraded first frame response. 
     To prevent first frame response degradation, Vth sampling should be performed after OBS 2  and prior to the emission phase.  FIG.  12    is a timing diagram illustrating how a voltage sampling and data programming operation can be inserted after the dominant on-bias stress phase OSB 2  to improve first frame response in accordance with another embodiment. As shown in  FIG.  12   , a first Vth sampling and data programming phase may be performed from time t 3 -t 4 , dominant on-bias stress phase OBS 2  may be performed from time t 5 -t 6 , and a second Vth sampling and data programming phase may be performed from time t 7 - 78  after OBS 2  and before the emission phase. As described above in connection with  FIGS.  3  and  11   , the brief period of time following the assertion of signal SC 2 ( n ) at time t 7  where the drive transistor is temporarily activated to charge gate voltage Vg up to (Vdata−Vth) represents a momentary on-bias stress phase OBS 3 . By the end of time t 8 , the source voltage Vs at Node 1  will be set to Vdata, whereas Vg and Vd will both be set to (Vdata−Vth) by virtue of the diode connection of the drive transistor. 
     Performing another Vth sampling and data programming operation after OBS 2  can help accommodate for any potential Vth drift during OBS 2 , thereby improving first frame response. Although the short on-bias stress phases such as OBS 1  and OBS 3  do occur during the refresh frame, the longer on-bias stress phase OBS 2  still dominates and if matched with OBS 2 ′ of the vertical planking frame, flicker can be minimized. Another potential issue that could arise to cause mismatch between OBS 2  and OBS 2 ′ is that the data signal applied to pixel  22  during the different periods might be different. As shown in the example of  FIG.  12   , Vsg across the drive transistor during OBS 2  could be (VH−(Vdata 1 −Vth)) while Vsg across the drive transistor during OBS 2  might be (VH−(Vdata 2 −Vth)), where Vdata 1  is not equal to Vdata 2 . If Vdata 1 ≠Vdata 2 , the on-bias stress voltage between the refresh frame and the vertical blanking frame will be different, which could still result in noticeable flicker and/or a low-gray optical response. 
     To compensate for any potential mismatch in data signals between OBS 2  and OBS 2 ′, the row-based initialization voltage Vdini(n) may be dynamically adjusted to slightly different voltage levels and/or the anode reset voltage Var may be dynamically tuned to slightly different voltage levels when transitioning between the refresh frame and the vertical blanking frame.  FIG.  13    is a timing diagram illustrating how voltages Vdini(n) and/or Var can be dynamically adjusted to match the on-bias stress during the refresh and the vertical blanking periods. As shown in  FIG.  13   , initialization voltage Vdini(n) may be raised from VL to VH during OBS 2  of the refresh frame but may be raised from VL to VH′ during OBS 2 ′ of the vertical blanking frame. The difference in VH (i.e., VH′−VH) may be equal to (Vdata 2 −Vdata 1 ) to help compensate for any mismatch in data signals, thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. 
     If desired, anode reset voltage Var may also be tuned to help reduce any mismatch between the refresh and the vertical blanking periods. As shown in  FIG.  13   , anode reset voltage Var may be tuned from a nominal voltage level Var_nom during the refresh frame to an adjusted voltage level Var_adj during the vertical blanking frame. The difference in Var (i.e., Var_adj-Var_nom) may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. In contrast to initialization voltage Vdini(n), which is a row-based signal, anode reset voltage Var may be a sub-frame based signal (e.g., Var need not be adjusted on a per-row basis but can be adjusted when switching from the refresh frame to the vertical blanking frame). The sub-frame based tuning of anode reset voltage Var may be performed by itself (without raising the initialization voltage to VH′) or in conjunction with raising the initialization voltage to VH′ to minimize undesired display artifacts and optimize display performance. 
       FIG.  14 A  is a diagram showing how a display pixel of the type shown in  FIG.  10    can be configured to support external current sensing operations.  FIG.  14 B  is a timing diagram showing the behavior of relevant row control signals to perform such external current sensing operations. As shown in  FIG.  14 B , scan control SC 3 ( n ) can be pulsed low to perform the initialization phase, and scan control signal SC 2 ( n ) can then be pulsed low to perform the Vth sampling and data programming phase. At some later time (e.g., when the display is turned off/idle or during some other time when the user is not viewing the display), both control signals SC 3 ( n ) and SC 2 ( n ) are simultaneously asserted while SC 1 ( n ) is deasserted to perform the current sensing operations. 
     Referring back to  FIG.  14 A , a low SC 1 ( n ) during the current sensing phase turns off transistor Toxide, whereas a low SC 3 ( n ) and a low SC 2 ( n ) will turn on transistors Tdata and Tini, respectively. Emission control signal EM(n) should be deasserted during this time, which deactivates transistors Tem 1  and Tem 2 . Configured in this way, a sensing current can flow from data line  310  through transistors Tdata, Tdrive, and Tini onto initialization line  308 ′, as indicated by sensing current path  1400 . The initialization voltage Vdini(n) should be set to the low voltage VL during current sensing operations. Current  1400  can be measured using sense circuitry  25  (see  FIG.  2   ) to generate compensation data that is stored in storage circuitry  29 . As described above, external Vth compensation via current sensing in combination with in-pixel Vth canceling can help minimize any undesired TFT effects associated with the threshold voltage of the drive transistor, which helps maintain consistent luminance levels over the lifespan of the display. 
     The example of  FIG.  13    in which three separate on-bias stress phases (e.g., OBS 1 , OB 2 , and OBS 3 ) are performed prior to the emission phase is merely illustrative.  FIG.  15    illustrates another suitable operational method where OBS 1  is removed. As shown in  FIG.  15   , only two separate on-bias stress phases (e.g., OBS 2  and OBS 3 ) are performed prior to the emission phase. Thus, scan control signal SC 2 ( n ) need to be pulsed only once during each refresh frame (i.e., during OBS 3 ). Note that by removing OBS 1 , the leading pulse in SC 3 ( n ) can also be removed (comparing  FIG.  13    to  FIG.  15   ), which allows OBS 2  to be performed immediately following the emission phase. Operating the display in this way obviates the need to perform OBS 2 , which can help save power and improve performance. The remainder of the display operation is similar to that of  FIG.  13    and need not be described again in detail. If desired, voltages Vdini(n) and Var may be dynamically adjusted to help compensate for any mismatch in data signals, thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. 
       FIG.  16    is a timing diagram showing another suitable way for operating display pixel circuit  22  of the type shown in  FIG.  10   . As shown in  FIG.  16   , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active data refresh period. During period Δt 2 , a pre on-bias stress phase (pre-OBS) may be performed by selectively pulsing signals SC 3 ( n ) and SC 3 ( n +1) and dynamically adjusting Vdini(n) from a low voltage level V INI_L  to a high voltage level V INI_H . Asserting signal SC 3 ( n ) will turn on transistor Tini to apply V INI_H  to the drain terminal of the drive transistor, whereas asserting signal SC 3 ( n +3) will turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 1  high while pulsing signals SC 3 ( n ) and SC 3 ( n +1) low. Voltage Vdini is back at the V INI_L  level. Driving signal SC 1  high will turn on n-channel semiconducting-oxide transistor Toxide. Driving signal SC 3 ( n ) low will turn on transistor Tini to apply V INI_L  to the drain terminal of the drive transistor, whereas driving signal SC 3 ( n +3) high will turn on transistor Tar to again perform anode reset on the OLED. 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2  low while signal SC 1  is still high and while signals SC 3 ( n ) and SC 3 ( n +1) remain deasserted. Driving signal SC 2  low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post on-bias stress phase (post-OBS) may be performed by selectively pulsing signals SC 3 ( n ) and SC 3 ( n +1) while Vdini(n) is adjusted to V INI_H . Asserting signal SC 3 ( n ) will turn on transistor Tini to again apply V INI_H  to the drain terminal of the drive transistor, whereas asserting signal SC 3 ( n +3) will turn on transistor Tar to again perform anode reset at the OLED. 
     If desired, anode reset voltage Var may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, V INI_H  and Var can be different between the active and blanking periods. If desired, V INI_H , Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS phase during Δt 2  and the post-OBS phase during Δt 5  in the active frame will add a first additional post-OBS phase during Δt 8  and a second additional post-OBS phase during Δt 9  in the blanking frame. During the blanking period, the pulsing of signal SC 3 ( n +1) will also serve to carry out at least three corresponding anode resets using the adjusted Var voltage. The driving scheme illustrated in connection with  FIG.  16    may (as an example) be implemented using four gate drivers on each side of the pixel array, where the SC 2 , EM, Vdini, and SC 3 ( n +1) signal drivers are formed on a first side of the array and where the SC 1 , SC 2 , SC 3 , Vdini signal drivers are formed on a second side of the array, which is equivalent to five total gate drivers. 
       FIG.  17 A  illustrates another suitable implementation of display pixel  22 . The pixel structure of  FIG.  17 A  is similar to the pixel structure of  FIG.  10   , except the initialization transistor Tini is now coupled to the gate of the drive transistor and an additional dedicated on-bias stress transistor Tobs is connected to the source terminal of the drive transistor. In particular, transistor Tini may be implemented as an n-channel semiconducting-oxide transistor with a first source-drain terminal connected to the gate terminal of the drive transistor, a gate terminal configured to receive a fourth scan control signal SC 4 , and a second source-drain terminal connected to an initialization line on which initialization voltage Vini is provided. Transistor Tobs may be implemented as a p-channel silicon transistor with a first source-drain terminal connected to the source terminal of the drive transistor, a gate terminal configured to receive the third scan control signal SC 3  (shared with transistor Tar), and a second source-drain terminal connected to an on-bias stress line on which on-bias stress voltage Vobs is provided. The on-bias stress voltage Vobs may be set to some predetermined or suitable voltage level that can be applied to the drive transistor such that the sampled Vth will correspond as close to the desired Vdata as possible. 
     The example of  FIG.  17 A  in which transistors Toxide and Tini are implemented using n-type semiconducting-oxide transistors while the remaining transistors are implemented using p-type silicon transistors is merely illustrative. If desired, transistors Toxide and Tini may alternatively be implemented as p-type semiconducting-oxide transistors; any one or more of the other transistors Tem 1 , Tem 2 , Tdrive, Tdata, Tar, and/or Tobs may be implemented as n-type or p-type semiconducting-oxide transistors or n-type silicon transistors; pixel  22  may include more than eight or less than eight transistors; pixel  22  may include more than one capacitor, etc. 
       FIG.  17 B  is a timing diagram illustrating the operation of pixel  22  shown in  FIG.  17 A . As shown in  FIG.  17 A , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active data refresh period. During period Δt 2 , a pre on-bias stress phase (pre-OBS) may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 4  high while the other signals are deasserted. Driving signal SC 4  high will turn on n-channel semiconducting-oxide transistor Tini to apply initialization voltage Vini to the gate terminal of the drive transistor. Signal SC 3  is high at this time, so no anode reset will be performed during Δt 3 . 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2  low while signal SC 1  is still high and while signals SC 3 ( n ) and SC 4  are deasserted. Driving signal SC 2  low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post on-bias stress phase (post-OBS) may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will again turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset at the OLED. 
     If desired, anode reset voltage Var and the on-bias stress voltage Vobs may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var and Vobs at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, Vobs and Var can be different between the active and blanking periods. If desired, Vobs, Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS/anode reset (AR) phase during Δt 2  and the post-OBS/anode reset phase during Δt 5  in the active frame will add a first additional post-OBS/AR phase during Δt 8  and a second additional post-OBS/AR phase during Δt 9  in the blanking frame. During the blanking period, the pulsing of signal SC 3  will serve to carry out at least two corresponding anode resets using the adjusted Var voltage. The driving scheme illustrated in connection with  FIG.  17 A /B may (as an example) be implemented using three gate drivers on each side of the pixel array, where the SC 1 , SC 2 , and EM signal drivers are formed on a first side of the array and where the SC 2 , SC 3 , an SC 4  signal drivers are formed on a second side of the array, which is equivalent to five total gate drivers. 
       FIG.  18 A  illustrates another suitable implementation of display pixel  22 . The pixel structure of  FIG.  18 A  is similar to the pixel structure of  FIG.  17 A , except the initialization transistor Tini is now controlled by SC 1 ( n −2), which is the SC 1  signal from two rows above. Similar to the example of  FIG.  17 A , transistor Tini of  FIG.  18 A  may be implemented as an re-channel semiconducting-oxide transistor. The example of  FIG.  18 A  in which transistors Toxide and Tini are implemented using n-type semiconducting-oxide transistors while the remaining transistors are implemented using p-type silicon transistors is merely illustrative. If desired, transistors Toxide and Tini may alternatively be implemented as p-type semiconducting-oxide transistors; any one or more of the other transistors Tem 1 , Tem 2 , Tdrive, Tdata, Tar, and/or Tobs may be implemented as n-type or p-type semiconducting-oxide transistors or n-type silicon transistors; pixel  22  may include more than eight or less than eight transistors; pixel  22  may include more than one capacitor, etc. 
       FIG.  18 B  is a timing diagram illustrating the operation of pixel  22  shown in  FIG.  18 A . As shown in  FIG.  18 A , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active period. During period Δt 2 , a pre-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 1 ( n −2) high while the other signals are deasserted. Driving signal SC 1 ( n −2) high will turn on n-channel semiconducting-oxide transistor Tini to apply initialization voltage Vini to the gate terminal of the drive transistor. Signal SC 3  is high at this time, so no anode reset will be performed during Δt 3 . 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2  low while signal SC 1 ( n ) is still high and while signals SC 3  and SC 1 ( n −2) are deasserted. Driving signal SC 2  low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will again turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset at the OLED. 
     If desired, anode reset voltage Var and the on-bias stress voltage Vobs may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var and Vobs at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, Vobs and Var can be different between the active and blanking periods. If desired, Vobs, Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS/AR phase during Δt 2  and the post-OBS/AR phase during Δt 5  in the active frame will add a first additional post-OBS/AR phase during Δt 8  and a second additional post-OBS/AR phase during Δt 9  in the blanking frame. During the blanking period, the pulsing of signal SC 3  will serve to carry out at least two corresponding anode resets using the adjusted Var voltage. The driving scheme illustrated in connection with  FIG.  18 A /B may (as an example) be implemented using three gate drivers on each side of the pixel array, where the SC 1 , SC 2 , and SC 3  signal drivers are formed on a first side of the array and where the SC 1 , SC 2 , and EM signal drivers are formed on a second side of the array, which is equivalent to only four total gate drivers. 
       FIG.  19 A  illustrates yet another suitable implementation of display pixel  22 . The pixel structure of  FIG.  19 A  is similar to the pixel structure of  FIG.  18 A , except the initialization transistor Tini is now coupled to the drain terminal of the drive transistor. In particular, transistor Tini may have a first source-drain terminal connected to the drain terminal of the drive transistor, a gate terminal configured to receive fourth scan control signal SC 4 , and a second source-drain terminal connected to an initialization line on which initialization voltage Vini is provided. Unlike the example of  FIG.  18 A , transistor Tini of  FIG.  19 A  may be implemented as a p-channel silicon transistor. The example of  FIG.  19 A  in which only transistor Toxide is implemented using an n-type semiconducting-oxide transistor while the remaining transistors are implemented using p-type silicon transistors is merely illustrative. If desired, transistor Toxide may alternatively be implemented as a p-type semiconducting-oxide transistor; transistor Tini may be implemented as an n-type or p-type semiconducting-oxide transistor; any one or more of the other transistors Tem 1 , Tem 2 , Tdrive, Tdata, Tar, and/or Tobs may be implemented as n-type or p-type semiconducting-oxide transistors or n-type silicon transistors; pixel  22  may include more than eight or less than eight transistors; pixel  22  may include more than one capacitor, etc. 
       FIG.  19 B  is a timing diagram illustrating the operation of pixel  22  shown in  FIG.  19 A . As shown in  FIG.  19 A , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active period. During period Δt 2 , a pre-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 4  low while signal SC 1  is high. Driving signal SC 4  low will turn on p-channel silicon transistor Tini to apply initialization voltage Vini to the drain terminal of the drive transistor. Signal SC 3  is high at this time, so no anode reset will be performed during Δt 3 . 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2  low while signal SC 1  is still high and while signals SC 3  and SC 4  are deasserted. Driving signal SC 2  low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on (since SC 1  is high) to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will again turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset at the OLED. 
     If desired, anode reset voltage Var and the on-bias stress voltage Vobs may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var and Vobs at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, Vobs and Var can be different between the active and blanking periods. If desired, Vobs, Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS/AR phase during Δt 2  and the post-OBS/AR phase during Δt 5  in the active frame will add a first additional post-OBS/AR phase during Δt 8  and a second additional post-OBS/AR phase during Δt 9  in the blanking frame. During the blanking period, the pulsing of signal SC 3  will serve to carry out at least two corresponding anode resets using the adjusted Var voltage. The driving scheme illustrated in connection with  FIG.  19 A /B may (as an example) be implemented using three gate drivers on each side of the pixel array, where the SC 2 , SC 3 , and SC 4  signal drivers are formed on a first side of the array and where the SC 1 , SC 2 , and EM signal drivers are formed on a second side of the array, which is equivalent to five total gate drivers. 
       FIG.  20 A  illustrates yet another suitable implementation of display pixel  22 . The pixel structure of  FIG.  20 A  is similar to the pixel structure of  FIG.  19 A , except the initialization transistor Tini is now controlled by SC 2 ( n −1), a SC 2  signal from one row above.  FIG.  20 B  is a timing diagram illustrating the operation of pixel  22  shown in  FIG.  20 A . As shown in  FIG.  20 A , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active period. During period Δt 2 , a pre-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 2 ( n −1) low while signal SC 1  is high. Driving signal SC 2 ( n −1) low will turn on p-channel silicon transistor Tini to apply initialization voltage Vini to the drain terminal of the drive transistor. Signal SC 3  is high at this time, so no anode reset will be performed during Δt 3 . 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2 ( n ) low while signal SC 1  is still high and while signals SC 3  and SC 2 ( n −1) are deasserted. Driving signal SC 2 ( n ) low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on (since SC 1  is high) to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will again turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset at the OLED. 
     If desired, anode reset voltage Var and the on-bias stress voltage Vobs may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var and Vobs at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, Vobs and Var can be different between the active and blanking periods. If desired, Vobs, Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS/AR phase during Δt 2  and the post-OBS/AR phase during t 5  in the active frame will add a first additional post-OBS/AR phase during Δt 8  and a second additional post-OBS/AR phase during Δt 9  in the blanking frame. The driving scheme illustrated in connection with  FIG.  20 A /B may (as an example) be implemented using three gate drivers on each side of the pixel array, where the SC 1 , SC 2 , and SC 3  signal drivers are formed on a first side of the array and where the SC 1 , SC 2 , and EM signal drivers are formed on a second side of the array, which is equivalent to four total gate drivers. 
       FIG.  21 A  illustrates yet another suitable implementation of display pixel  22 . The pixel structure of  FIG.  21 A  is similar to the pixel structure of  FIG.  19 A , except the initialization transistor Tini is now controlled by SC 3 ( n −7), a SC 3  signal from seven rows above.  FIG.  21 B  is a timing diagram illustrating the operation of pixel  22  shown in  FIG.  21 A . As shown in  FIG.  21 A , the operations prior to time t 6  are performed during the active/refresh period, whereas the operations after time t 6  are performed during the blanking period. At time t 1 , the emission signal EM may be deasserted (e.g., driven high) to begin the active period. During period Δt 2 , a pre-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset for the OLED. 
     During period Δt 3 , an initialization phase may be carried out by pulsing signal SC 3 ( n −7) low while signal SC 1  is high. Driving signal SC 3 ( n −7) low will turn on p-channel silicon transistor Tini to apply initialization voltage Vini to the drain terminal of the drive transistor. Signal SC 3  is high at this time, so no anode reset will be performed during Δt 3 . 
     During period Δt 4 , a data programming/sampling phase may be performed by pulsing signal SC 2 ( n ) low while signal SC 1  is still high and while signals SC 3  and SC 3 ( n −7) are deasserted. Driving signal SC 2 ( n ) low will turn on transistor Tdata to load the desired data signal onto the source terminal of the drive transistor while transistor Toxide is kept on (since SC 1  is high) to allow sampling of the threshold voltage Vth of the drive transistor. 
     During period Δt 5 , a post-OBS/AR phase may be performed by selectively pulsing signal SC 3 ( n ). Asserting signal SC 3 ( n ) will again turn on transistor Tobs to apply Vobs to the source terminal of the drive transistor and will also turn on transistor Tar to perform anode reset at the OLED. 
     If desired, anode reset voltage Var and the on-bias stress voltage Vobs may be adjusted at time t 6  to help reduce any mismatch between the active and blanking periods. The voltage change in Var and Vobs at time t 6  may be any suitable voltage delta to help compensate for any operational mismatch within pixel  22 , thereby eliminating any residual flicker and closing any undesired luminance gaps when toggling between refresh and vertical blanking frames. At time t 7 , the emission signal EM may be deasserted (e.g., driven high) to begin the blanking period. During the blanking period, the data signals on the data line may be parked at some predetermined voltage level Vpark to help reduce dynamic power consumption. If desired, Vobs and Var can be different between the active and blanking periods. If desired, Vobs, Var, and Vpark may also be different between different blanking periods. 
     Note that the pre-OBS/AR phase during Δt 2  and the post-OBS/AR phase during Δt 5  in the active frame will add a first additional post-OBS/AR phase during Δt 8  and a second additional post-OBS/AR phase during Δt 9  in the blanking frame. The driving scheme illustrated in connection with  FIG.  21 A /B may (as an example) be implemented using three gate drivers on each side of the pixel array, where the SC 1 , SC 2 , and SC 3  signal drivers are formed on a first side of the array and where the SC 1 , SC 2 , and EM signal drivers are formed on a second side of the array, which is equivalent to four total gate drivers. 
     The embodiments of  FIGS.  1 - 21    describing display pixels  22  operable to support in-pixel Vth canceling and external current sensing are merely illustrative and are not intended to limit the scope of the present embodiments. In general, pixels  22  may be modified to include more than seven or less than seven thin-film transistors and can include more or less capacitors. The polarity of any of the pixel transistors can be flipped (e.g., p-type transistors can instead be implemented using n-type transistors and vice versa). More than one emission control signal can be used per row (e.g., transistor Tem 1  might be controlled using a first emission signal EM 1 , whereas transistor Tem 2  might be controlled using a second separate emission signal EM 2 ). More than two or less than two scan control signal can be used per row, where each scan control signal can optionally be shared among two or more neighboring rows of display pixels. If desired, other ways of implementing on-bias stress in high refresh rate or low refresh rate display can be adopted to mitigate the impact of hysteresis and minimize flicker. 
     The foregoing is merely illustrative and various modifications can be made by those skilled in the art without departing from the scope and spirit of the described embodiments. The foregoing embodiments may be implemented individually or in any combination.