Patent Publication Number: US-6222327-B1

Title: Lighting device for illumination and lamp provided with the same

Description:
REFERENCE TO EARLIER FILED APPLICATION(S) 
     This application is a continuation of the following earlier filed application(s): Ser. No. 09/096,453 filed Jun. 11, 1998, now U.S. Pat. No. 6,124,680; Ser. No. 08/921,363 filed Aug. 29, 1997, issued as U.S. Pat. No. 5,977,725, the disclosure of which is hereby incorporated by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates to a lighting device for a discharge tube. 
     In recent days, a discharge tube (for example, a fluorescent lamp) is more likely to employ a system arranged to convert a DC voltage into a high-frequency AC voltage through the effect of a lighting circuit composed of an inverter and to apply the high-frequency AC voltage into a resonant load circuit containing the discharge tube itself. The resonant load circuit also contains a resonant inductor and a resonant capacitor for setting a resonant frequency. This type of lighting circuit includes an inverter circuit composed of two power semiconductor switching elements connected between a positive and a negative polarities in a half-bridge structure, which is operated to apply the high-frequency AC voltage onto both ends of the resonant load circuit. The waveform of the current flowing through the resonant load circuit (referred to as resonant current) is resonated into a sinusoidal waveform through the effect of the inductor and the capacitor. This resonant current is controlled by changing an operation frequency of the inverter. 
     As a prior art for stabilizing the driving frequency of a switching element, there has been proposed a stabilizing circuit disclosed in JP-A-8-45685. This circuit provides a half-bridge circuit for supplying an AC voltage to the resonant load circuit containing the discharge lamp and is operated to divide part of the resonant current into a capacitor and a feedback transformer and apply a control signal to the switching element between a high side and a low side of the half-bridge circuit according to a voltage on the secondary side of the feedback transformer. Unlike the normal fluorescent lamp, this prior art discusses the lighting device for an electrodeless fluorescent lamp for generating plasma through magnetic lines of force emitted from an exciting coil. The electrodeless lamp operates to supply high-frequency current of several megahertz to a solenoid type exciting coil and thereby emit magnetic lines of force for generating ions inside of a bulb through the inductive discharge and form those ions as discharge current in a closed loop (plasma) through the effect of electromagnetic coupling applied by the force of magnetization. Mercury vapor in the plasma is excited by the inductive electric field so as to fire ultraviolet rays onto a fluorescent material coated on the inside of the tube on which the ultraviolet rays are converted into visible rays. 
     SUMMARY OF THE INVENTION 
     The foregoing prior art is arranged to supply a control signal having the same frequency as the resonant current to the half-bridge circuit through the feedback transformer. That is, this prior art is a self-excited circuit that keeps the half-bridge circuit in operation without any signal supplied from the external and thus is suitable to the high-frequency operation. However, the feedback transformer has self-inductance, which brings about a phase difference between the control signal and the resonant current and furthermore a slip of the frequency of the control signal from a suitable value. The phase difference and the slip of the frequency may allow pass current to flow through the half-bridge circuit, thereby disadvantageously increasing the loss. Hence, a first problem to be solved by the present invention is to provide lighting circuit means which stabilizes at high frequency. 
     The electrodeless fluorescent lamp to be controlled by the prior art provides an exciting coil. The use of the exciting coil as a resonant inductor may be effective in reducing the cost of parts and the circuit in size. However, the plasma may given an influence onto an equivalent inductance of the exciting coil, which means that the equivalent inductance is varied depending on the luminescent state of the lamp. Hence, a second problem to be solved by the present invention is to implement the lighting circuit that may use the exciting coil of the electrodeless fluorescent lamp as the resonant inductor. 
     As described above, an object of the present invention is to provide a lighting device which stably operates at a high frequency and is small in size and cost. 
     The foregoing object may be achieved by a lighting device for illumination for applying an AC voltage to resonating means having an inductor and a capacitor according to the switching operation of two power semiconductor elements connected in bridge and supplying AC current to a discharge tube connected to any one of the inductor and the capacitor, comprising first and second voltage drop means and said resonating means connected in series with an I/O terminal of the bridge, the voltages of the first and the second voltage drop means being applied to each control terminal of the two power semiconductor elements as a signal of an opposite phase. 
     The object of utilizing the exciting coil of the electrodeless fluorescent lamp may be achieved by comprising a bridge circuit having two power semiconductor elements, a capacitor, the exciting coil, and the first and the second voltage drop means connected in series between I/O terminals of the bridge circuit, wherein the voltages of the first and the second voltage drop means may be applied to each control terminal of the two power semiconductor elements as a signal of an opposite phase through first and second phase shift means. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a first schematic view showing a lighting circuit according to the present invention. 
     FIG. 2 is a second schematic view showing a lighting circuit according to the present invention. 
     FIG. 3 is a view showing a lighting circuit according to a first embodiment of the present invention. 
     FIG. 4 is a view showing a first equivalent circuit of a gate circuit shown in FIG.  3 . 
     FIG. 5 is a view showing a second equivalent circuit of a gate circuit shown in FIG.  3 . 
     FIG. 6 is an explanatory view showing an operation of an embodiment shown in FIG.  3 . 
     FIG. 7 is a graph showing relation between a phase difference between a resonant current and a gate voltage to be applied to a gate inductor and an operating frequency. 
     FIG. 8 is a view showing a lighting circuit according to a second embodiment of the present invention. 
     FIG. 9 is an explanatory view showing an operation of the embodiment shown in FIG.  8 . 
     FIG. 10 is a third schematic diagram showing a lighting circuit according to the present invention. 
     FIG. 11 is a view showing an equivalent circuit of an electrodeless lamp. 
     FIG. 12 is a view showing an equivalent circuit of an electrodeless lamp before lighting. 
     FIG. 13 is a view showing an equivalent circuit of an electrodeless lamp after lighting. 
     FIG. 14 is a graph showing relation between frequency and resonant current. 
     FIG. 15 is a view showing a lighting circuit according to a third embodiment of the present invention. 
     FIG. 16 is a graph showing relation between a resonant load frequency and a phase difference between resonant current and a gate voltage. 
     FIG. 17 is a graph showing waveforms of a drain voltage Vds, a gate voltage Vg and a resonant current IL of a gate circuit. 
     FIG. 18 is a view showing a first embodiment of a gate circuit. 
     FIG. 19 is a view showing a second embodiment of a gate circuit. 
     FIG. 20 is a view showing a lighting circuit according to a fourth embodiment of the present invention. 
     FIG. 21 is a view showing a lighting circuit according to a fifth embodiment of the present invention. 
     FIG. 22 is an explanatory view showing an operation of the embodiment shown in FIG.  21 . 
     FIG. 23 is a view showing a lighting circuit according to a sixth embodiment of the present invention. 
     FIG. 24 is an explanatory view showing an operation of the embodiment shown in FIG.  22 . 
     FIG. 25 is a view showing a lighting circuit according to a seventh embodiment of the present invention. 
     FIG. 26 is a view showing a lighting circuit according to an eighth embodiment of the present invention. 
     FIG. 27 is a fourth schematic view showing a lighting circuit according to the present invention. 
     FIG. 28 is a fourth schematic view showing a lighting circuit according to the present invention. 
     FIG. 29 is a sixth schematic view showing a lighting circuit according to the present invention. 
     FIG. 30 is a view showing a lighting circuit according to a ninth embodiment of the present invention. 
     FIG. 31 is an explanatory view showing an operation of the embodiment shown in FIG.  30 . 
     FIG. 32 is a view showing a lighting circuit according to a tenth embodiment of the present invention. 
     FIG. 33 is an explanatory view showing an operation of the embodiment shown in FIG.  32 . 
     FIG. 34 is a view showing a lighting circuit according to an eleventh embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     FIG. 1 is a lighting circuit for supplying AC current to a discharge tube  16 . The discharge tube  16  is mainly intended for an electrodeless fluorescent lamp. A voltage source  15  operates to supply a DC voltage to the lighting circuit of the discharge tube  16 . Normally, the voltage source  15  operates to rectify an AC voltage through a diode-bridged rectifying circuit and generate a DC voltage. A pair of switching elements Q 1  and Q 2  connected in series is connected between a positive and a negative electrodes of the voltage source  15 , wherein a contact between the switching elements is denoted by O and a contact between the switching element Q 2  and the negative electrode of the voltage source  15  is denoted by N. Between the O point and the N point is series-connected voltage drop means Z 1  and Z 2 , a resonant inductor Lr and a resonant capacitor Cr, in which the capacitor Cr provides a discharge tube (or fluorescent lamp) as a load in parallel. Those components compose a resonant load circuit, the current frequency of which is determined by the value of each component. The mutual switching operation of the switching elements Q 1  and Q 2  is executed to flow bidirectional current through the resonant load circuit, thereby lighting up the discharge tube. The switching element Q 1  or Q 2  is an n-channel MOSFET, for example, which provides a drain terminal to which current is input, a source terminal from which current is output, and a gate terminal to and from which a control voltage is applied or removed. The application or the removal of the control voltage to or from the gate terminal allows the current flowing between the drain and the source to be conducted or cut off. The MOSFET contains a built-in diode located in the direction from the source terminal to the drain terminal. The diode built in Q 1  is QD 1  and the diode built in Q 2  is QD 2 . In FIG. 1, the gate circuits of the switching elements Q 1  and Q 2  correspond to the voltage drop means Z 1  and Z 2  of the resonant load circuit. The voltage generated by flowing current of the resonant load circuit through Z 1  and Z 2  is applied to the gate for the purpose of controlling the switching operation of the switching elements Q 1  and Q 2 . The voltages of the voltage drop means Z 1  and Z 2  caused by the current of the resonant load circuit are opposite in polarity to each other with the O point or the N point as a reference level, which causes the switching elements Q 1  and Q 2  to be alternately switched. This keeps the self-excited drive in synchronous to the frequency of the current of the resonant load current. 
     In the circuit of FIG. 1, the on and off timing of the switching elements Q 1  and Q 2  is determined by the voltage drop means Z 1  and Z 2 . For adjusting brightness of the discharge tube, the values of Z 1  and Z 2  are selected. Herein, the adjustment of the brightness of the discharge tube may be achieved by changing the magnitude of resonant current IL. As a switching frequency of the lighting circuit is rising against a resonant frequency of determined by the resonant inductor and the resonant capacitor, the current IL is decreasing. Based on this principle, the lighting device is operated to tune light by controlling the switching frequency f. For example, for reducing the current IL, it is possible to shorten the conducting period of the switching element and thereby enhance the switching frequency. As stated above, in FIG. 1, the voltage drop means operates to tune brightness of the discharge tube. FIG. 2 shows another lighting circuit provided with means for turning brightness. The similar parts of FIG. 2 to those of FIG. 1 have been described with respect to FIG.  1 . Hence, the description about those parts is left out. In FIG. 1, the gate circuits of the switching elements Q 1  and Q 2  are the voltage drop means Z 1  and Z 2  of the resonant load circuit, while in FIG. 2, phase shift means Z 3  and Z 4  are provided between the gate terminals of the switching elements Q 1  and Q 2  and the voltage drop means Z 1  and Z 2 . The phase shift means serve to delay or advance a phase when the voltage of the voltage drop means is applied between a gate and a source of Q 1  or Q 2 . This type of method makes it possible to optionally adjust the on and off timing of Q 1  and Q 2 , thereby being able to tune light. 
     In FIG. 2, the voltage drop means Z 1 , Z 2  or the phase shift means Z 3 , Z 4  may be a capacitor, an inductor, a resistor or a combination of those components, for example. FIG. 3 shows an embodiment of a concrete lighting circuit having passive elements as the voltage drop means Z 1 , Z 2  and the phase shift means Z 3 , Z 4 . In this embodiment, the capacitor C 1  served as the voltage drop means is provided between the contact O and the resonant inductor Lr and the capacitor C 2  is provided between the resonant capacitor Cr and the contact N. The C 1  is connected in parallel to a resistor R 1  and the C 2  is connected in parallel to a resistor R 2 . As the resistance is selectively made smaller with the DC components to be overlapped with the voltages of C 1  and C 2 , the voltage of the capacitor is made more variable in similar amplitude with the zero voltage as a reference. The inductor L 1  and the resistor Rd 1  both served as phase shift means are connected in series between one end of the capacitor C 1  and one gate end of Q 1 . L 1  and Rd 1  bring about a phase lag when a voltage of C 1  is applied between the gate and the source of Q 1 . Likewise, L 2  and Rd 2  bring about a phase lag when a voltage of C 2  is applied between the gate and the source of Q 2 . Zener diodes ZD 7  and ZD 8  coupled in series and in opposite are provided in parallel between the gate and the source of Q 1 . Likewise, Zener diodes ZD 7  and ZD 8  are provided in parallel between the gate and the source of Q 2 . When an excessive voltage is applied between the gate and the source, those Zener diodes serve to prevent breakage of the switching elements. The MOSFET having a Zener diode for preventing an excessive voltage built therein has been made commercially available. In the case of selecting this type of switching element, the foregoing Zener diodes may be removed from the circuit arrangement. 
     In FIG. 3, for enabling Q 1  and Q 2  to be alternately switched, it is necessary to provide a circuit for starting the switching operation. Then, the starting circuit will be described below. The starting circuit is arranged to connect resistors Rs 1 , Rs 2  and a starting capacitor Cs in series between a positive and a negative electrodes of the voltage source  15 , in which a contact between Rs 1  and Rs 2  is connected to a contact O between Q 1  and Q 2 . The capacitor Cs is charged up to a starting voltage by the voltage source  15  that feeds a voltage to the capacitor Cs through the resistors Rs 1  and Rs 2 . Between a gate terminal of Q 2  and a contact between Rs 2  and Cs is provided a bi-directional thyristor  17  called “SIDAC” (Silicon Diode for Alternating Current), which is served as a breakover voltage type switch. In this starting circuit, if the voltage of the starting capacitor Cs is equal to or lower than the breakover voltage of SIDAC  17 , the SIDAC  17  is off, when the resonant capacitor Cr is charged by the voltage source  15  through the resistor Rs 1 . When the voltage of the starting capacitor Cs reaches the breakover voltage, the SIDAC  17  changes the off state into an on state, in which the charges stored in Cs are moved to the capacitance between the gate and the source of Q 2 . As a result, the switching element Q 2  is turned on so that the voltage charged in the resonant capacitor Cr allows the current to flow through the resonant load circuit, thereby enabling the switching elements Q 1  and Q 2  to alternately start the switching operation. In the constant operating state, for suppressing the action of the starting circuit, it is necessary to suppress the voltage of the starting capacitor Cs to the breakover voltage or lower of the SIDAC  17 . In the constant operating state, a voltage at the contact O between Q 1  and Q 2  corresponds to an alternate voltage of the positive and the negative electrodes of the voltage source  15 . Hence, the time constant of the resistor Rs 2  and the capacitor Cs may be set so that the voltage of Cs is lower to the breakover voltage of the SIDAC  17  or lower. 
     In turn, the gate circuit will be described in detail. For this description, the gate circuit on a high voltage side is used. If the gate circuits are equivalently expressed, each gate circuit may be distinguished on the state of the Zener diode connected to each switching element. FIG. 4 shows an equivalent circuit provided when the Zener diode is turned on. Letting Rz denote an internal resistor of the Zener diode, the resistor Rz is connected in series with the phase shift means L 1  and Rd 1 . In FIG. 4, a synthetic impedance Zg of the resistor Rz and the phase shift means Rd 1  and L 1  is inductive. In this case, the current ig flowing through L 1 , Rd 1 , and Rz indicates a lagged phase rather than the voltage Vc 1  of the capacitor with the contact O as a reference level. On the other hand, when the Zener diode is off, the equivalent circuit corresponds to the circuit shown in FIG.  5 . Letting Ciss be an input capacitance of Q 1 , Ciss is connected in series with the phase shift means  11  and Rd 1 . In FIG. 5, the synthetic impedance Zg of Ciss, L 1  and Rd 1  is capacitive or inductive depending on the relation of the magnitude and the frequency between Ciss and L 1 , or if the reactance of Ciss has the same value as that of L 1 , the synthetic impedance is made to correspond to the resistance. Hence, the current ig flowing the impedance Zg composed of Ciss, L 1  and Rd 1  indicates an advance or lagged phase or same phase rather than the voltage Vc 1  of C 1  with the contact O as a reference level. 
     Herein, in the lighting circuit as shown in FIG. 3, the maximum value of the current flowing through the resonant load circuit is variable depending on the lighting state of the discharge tube  16 . When the discharge tube is not lit, the current flowing through the circuit is made larger, so that the voltages Vc 1  and Vc 2  of the capacitors C 1  and C 2  are increased. Letting Vz be a Zener voltage of the Zener diode connected to Q 1 , if Vc 1  exceeds Vz, the Zener diode is in the on state, in which case the gate circuit is made to correspond to the equivalent circuit shown in FIG.  4 . After the discharge tube is lit, the current flowing through the circuit is reduced, so that the voltages Vc 1  and Vc 2  of the capacitors C 1  and C 2  are reduced accordingly. If Vc 1  and Vc 2  are equal to or lower than Vz, the gate circuit is made to correspond to the equivalent circuit shown in FIG.  5 . Herein, the voltage of Ciss is equal to the voltage applied between the gate and the source of Q 1 . The voltage has a waveform lagged by π/2 [rad] rather than the current flowing through the pass of L 1 , Rd 1  and Ciss. Letting iL be current flowing through the resonant load circuit, vc denote a voltage of C 1 , and vg denote a voltage between the gate and the source, il, vc and vg may be represented as follows.              iL   =     Im                 sin                 ω                 t                 vc   =     Im           Z             sin        (       ω                 t     +     φ                 z       )                     vg   =           Im           Z               Zg          ·     1     ω                 Ciss              sin        (       ω                 t     +     φ                 g       )                               
     Herein, letting Z be a synthetic impedance of the circuit shown in FIG.  5  and Zg be impedance of Ciss, Li and Rd 1  of the circuit shown in FIG. 5, φz denote a phase difference between iL and vc, and φg denote a phase difference between iL and vg. φz is variable depending on the impedances of the voltage drop means Z 1 , Z 2  and the phase shift means Z 3 , Z 4 . As stated above, φg may be a positive value or a negative one depending on the characteristic of the impedance Zg composed of Ciss, L 1  and Rd 1  of the gate circuit. 
     Next, the operation of the circuit shown in FIG. 3 will be described with reference to FIG.  6 . FIG. 6 represents a waveform of each component in the embodiment shown in FIG.  3 . The discharge tube  16  is provided with high-frequency current by the resonant load circuit composed of Q 1 , Q 2  and Lr, Cr. Assuming that the direction of flowing the current IL of the resonant load circuit from the O point in FIG. 3 is defined to be positive, during one period of the current IL, four operation modes are provided about Q 1 , Q 2  and QD 1 , QD 2 . These periods are indicated as t 1  to t 4  in FIG.  6 . Later, each operation mode will be described. 
     Mode 1 (t 1  period): When the switching element Q 1  is turned on, the current IL is flown from the capacitor  15  through the passage of Q 1 , C 1 , Lr, Cr and C 2 . The current IL is charged in the capacitor Cr and part of the current IL is branched into the discharge tube  16 . Though the current IL is charged in the capacitor C 1 , the voltage of C 1  is represented as Vc 1 . At the mode 1, the voltage applied between the gate and the source of the switching element Q 1  has a voltage waveform depicted in a real line, which lags behind the voltage Vc 1  in phase. This voltage waveform indicates a long time passed until the gate voltage of the switching element Q 1  becomes lower than a threshold value, that is, until the switching element Q 1  is turned off. When the gate voltage is equal to or lower than the threshold voltage of the MOSFET, Q 1  is turned off. 
     On the other hand, the capacitor C 2  is charged by IL. Letting Vc 2  be a voltage of C 2 , the capacitor voltage Vc 2  indicated in a broken line with an N point as a reference level is increased. The voltage applied between the gate and the source of Q 2  is made to have a voltage waveform indicated in a real line, which lags behind the voltage Vc 2  in phase. This voltage waveform indicates a long time passed until the gate voltage of Q 2  is made lower than the threshold value, that is, until the gate voltage of Q 2  is turned off. The present system arranged to make the current IL have a sinusoidal waveform through the effect of Lr and Cr and to turn off Q 1  according to the voltage of Vc 1  is characterized in turning off Q 1  while the current IL keeps its polarity positive. If the value of the voltage drop means C 1  is equal to the value of the voltage drop means C 2 , the voltages Vc 1  and Vc 2  caused by the current IL flowing through C 1  and C 2  are equal in magnitude to each other but opposite in polarity to each other with the O point and the N point as a reference level. 
     Mode 2 (t 2  period): When Q 1  is turned off, the current IL has a value in positive polarity. This current is kept flowing through the passage of Lr, Cr, C 2 , QD 2  and C 1 . In addition, part of the current IL is branched into the discharge tube  16 . 
     The current IL serves to charge the capacitor C 2 . Vc 2  is increased and the gate voltage of Q 2  is increased with the N point as a reference level. When the gate voltage is greater than or equal to the threshold voltage of the MOSFET, Q 2  is turned on. Further, during the mode 2 period, the current polarity is opposite in polarity with Q 2  as a reference. As long as the current polarity is kept stable if the gate voltage is charged as shown in FIG. 6, the current is kept flowing through QD 2 . The mode 2 indicates the period when the polarity of the current IL is changed from the positive to the negative. During this mode period, the voltage Vc 1  of C 1  is further reduced. This allows the voltage of Vc 1  to be applied in a reversely biased manner. Hence, the gate voltage of Q 1  does not cause Q to be instaneously turnned on again so that Q 1  may be stably turned off. 
     Mode 3 (t 3  period): When the current IL changes from a positive polarity to a negative one, the current IL is flown in Q 2  charged with the gate voltage at the mode 2. That is, the current IL is flown as the discharge current of Cr through a passage of Q 2 , C 2 , Cr, Lr and C 1 , and then C 2  is charged with the current IL. The current IL serves to reduce the voltage Vc 2  so that the voltage Vc 2  is made equal to or lower than the threshold voltage of the MOSFET, the switching element Q 2  is turned off. Also at the mode 3, like the mode 1, while the current IL keeps the negative polarity, Q 2  is turned off. On the other hand, the voltage of C 1  is increased with the O point as a reference level. 
     Mode 4 (t 4  period): When Q 2  is turned off, the current IL has a value in negative polarity. The current IL is kept flowing through a passage of Lr, C 1 , Q 1 , the voltage source  15 , C 2 , Cr, and Lr. In addition, part of the current IL is branched into the discharge tube  16 . 
     The current IL is charged in C 1 . With increase of Vc 1 , the gate voltage of Q 1  is increased. When the gate voltage is greater than or equal to the threshold voltage of the MOSFET, Q 1  is turned on. However, during the mode 4 period, the polarity of the current is opposite as viewed from Q 1 . As long as the polarity of the current is kept stable if the gate voltage is charged, the current is kept flowing in QD 1 . The mode 4 corresponds to the period when the polarity of the current IL is changed to a positive value. During this period, the voltage Vc 2  of C 2  is reduced. 
     As described above, during one period of the current IL, the operation from the mode 1 to the mode 4 is executed. Later, this operation is repeated. 
     In FIG. 3, letting φg be a phase difference of a voltage between the gate and the source of Q 1  and Q 2  and fs be an operating frequency of the lighting circuit with the values of gate inductors L 1  and L 2  served as phase shift means as parameters, the characteristic is made to be that shown in FIG.  7 . The characteristic will be described with reference to FIG.  5 . In the series circuit composed of Ciss, Rd 1  and L 1 , if the reactance of Ciss is larger than that of L 1 , that is, the reactance of Ciss is capacitive, with increase of L 1 , the impedance of the series circuit is coming closer to an inductive value. The resulting reactance delays the current flowing through the series circuit and the gate voltage corresponding to the voltage of Ciss. Hence, since the phase difference φg of the gate voltage against the resonant load current is made smaller and the conducting period of the switching element is made longer, the switching frequency is made lower. As mentioned above, the provision of the phase shift means makes it possible to optionally adjust the on and off timing of Q 1  and Q 2 , thereby being able to change the operating frequency. 
     The foregoing embodiments have been arranged to apply a voltage of the voltage drop means to the switching elements for enabling the switching elements to alternately do the switching operation. On the other hand, the lighting circuit arranged to use the voltage drop means as means for detecting resonant current is illustrated in FIG.  8 . The arrangement of the resonant load circuit is likewise to that shown in FIG.  1 . Hence, the description thereabout is left out. The driving circuit  11  on the high side for driving Q 1  will be described with reference to FIG.  8 . The driving circuit  11  provides as a power source a capacitor  13  having a contact O between Q 1  and Q 2  as a reference level. By turning on Q 2 , the voltage of a capacitor C 14  having an N point as a reference level is charged in a diode D 1 . This method is called a Bootstrap system, which is described in U.S. Pat. No. 4,316,243. A CMOS inverter composed of elements  1  and  2  is provided between a positive and a negative electrodes of the capacitor  13  and serves to feed its output to the gate of Q 1 . When the element  1  is turned on (at this time the element  2  is off), the CMOS inverter operates to flow the current with which the voltage is applied to the gate terminal of Q 1 . Then, when the element  2  is turned on (at this time the element  1  is off), the CMOS inverter serves to flow the current with which the charges charged in the gate terminal of Q 1  are discharged. A NAND circuit  5  operates to feed a signal to a control terminal of the CMOS inverter composed of the elements  1  and  2 . The voltage of the capacitor C 1  is compared with a reference voltage Vrefl with the contact O as a reference level through the effect of a comparator  6 . The output of the comparator  6  is applied into the NAND circuit  5 . The comparator  16  is inputted with a positive voltage by a power source  15 . Further, a starting and stopping means connected in series to the resistor R 3  and the switch S 1  is provided between both terminals of the capacitor  13 . The contact between R 3  and S 1  is connected to an input of the NAND circuit  5 . In FIG. 8, the lighting circuit is started by turning off S 1 , while it is stopped by turning on S 1 . 
     In turn, the description will be oriented to the driving circuit  12  on the low side. The driving circuit  12  has the same arrangement as the driving circuit  11  on the hight side. A power source  12  indicates a capacitor with the N point as a reference level. Between a positive and a negative electrodes of the capacitor  14  is provided a CMOS inverter composed of elements  3  and  4 , the output of which is connected to the gate of Q 2 . The CMOS inverter composed of the elements  3  and  4  provides a control terminal input with a signal from the NAND circuit  7 . The voltage of the capacitor C 2  is compared with a reference voltage Vref 2  with the N point as a reference level through the effect of the comparator  8 . The output of the comparator  8  is applied into the NAND circuit  7 . It is desirable that the reference voltage Vref 1  on the high side is equal to the reference voltage Vref 2 . Between both terminals of the capacitor  14  is provided a starting and stopping means connected in series to the resistor R 4  and the switch S 2 . The contact between R 4  and S 2  is connected to an input of the NAND circuit  7 . Like S 1 , by turning off S 2 , the lighting circuit is started, while by turning of S 2 , the lighting circuit is stopped. 
     In turn, the description will be oriented to the operation of the lighting circuit with reference to FIG.  9 . FIG. 9 shows a waveform of each component included in the embodiment shown in FIG.  8 . Later, each operation mode will be described with reference to FIG.  9 . 
     Mode 1 (t 1  period): When Q 1  is turned on, the voltage source  15  starts to flow the current IL through a passage of Q 1 , C 1 , Lr, Cr and C 2 . The current IL is charged in the capacitor C 1  and the voltage Vc 1  is reduced with the O point as a reference level. The voltage Vc 1  is compared with a reference voltage Vref 1  (VHL) through the effect of the comparator  6 . When the voltage Vc 1  is made lower than the reference voltage Vref 1 , the output of the comparator  6  is changed from HIGH to LOW. This output is received by the NAND circuit  5 . Then, the element  2  of the CMOS inverter is turned on so that Q 1  discharges the gate voltage and is turned off. 
     The operation continued up to this point is the mode 1. The capacitor C 2  is charged by the current IL, so that the voltage Vc 2  is increased but does not reach the reference voltage Vref 2  (VLH). Hence, Q 2  is kept off. 
     Mode 2 (t 2  period): When Q 1  is turned off, the current IL has a value of a positive polarity, so that the current is kept flowing through a passage of Lr, Cr, C 2 , QD 2  and C 1 . In addition, part of the current IL is branched into the discharge tube  16 . 
     The current IL is charged in C 2  so that Vc 2  is increased with the N point as a reference level. When Vc 2  reaches Vref 2  (VLH), the output of the comparator  8  is changed from LOW to HIGH and is received by the NAND circuit  7 . The element  3  of the CMOS inverter is turned on for charging the gate voltage of Q 2 . During the mode 2 period, the current polarity is opposite as viewed from Q 2 . As shown in FIG. 8, if the gate voltage is charged, the current is kept flowing through QD 2  as long as the current polarity is changed. The period to change of the polarity of the current IL to a negative one corresponds to the mode 2, during which the voltage Vc 2  of C 2  is progressively increased and the voltage Vc 1  of C 1  is further reduced. 
     Mode 3 (t 3  period): When the polarity of the current IL is changed from a positive value to a negative one, the current IL starts to flow through Q 2  whose gate voltage is charged at the mode 2. That is, the current IL flows as discharge current through Q 2 , C 2 , Cr, Lr, and C 1 . The current IL serves to reduce the voltage Vc 2 . The Vc 2  is compared with the Vref 2  (VHL) by the comparator  8 . When the Vc 2  is lower than the Vref 2 , the output of the comparator is changed from HIGH to LOW and then is received by the NAND circuit  7 . Then, the element  4  of the CMOS inverter is turned on so that the Q 2  discharges the gate voltage of Q 2  and then is turned off. 
     The operation up to this point corresponds to the mode 3. The capacitor C 1  is charged by the current IL. The Vc 1  is increased but does not reach the reference voltage Vref 1  (VLH). Hence, the Q 1  is kept off. 
     Mode 4 (t 4  period): When the Q 2  is turned off, the current IL has a value of a negative polarity. The electromagnetic energy accumulated in Lr serves to keep the current IL flowing through a passage of Lr, C 1 , QD 1 , the voltage source  15 , C 2 , Cr, and Lr. In addition, part of the current IL is branched into the discharge tube  16 . 
     The current IL is charged in C 1  so that Vc 1  is increased. When Vc 1  exceeds Vref 1  (VLH), the output of the comparator  6  is changed from LOW to High and then is received by the NAND circuit  5 . Then, the element  1  of the CMOS inverter is turned on so that the gate voltage of Q 1  is charged. However, during the mode 4 period, the current polarity is opposite as viewed from Q 1 . Hence, if the gate voltage is charged, the current is kept flowing through QD 1  as long as the current polarity is changed. The period to the change of the polarity of the current IL to a positive value corresponds to the mode 4. During this period, the voltage Vc 1  of C 1  is progressively increased and the voltage Vc 2  of C 2  is further reduced. 
     During one period of the current IL, the operation from the mode 1 to the mode 4 is executed. After that, this operation is repeated. 
     In turn, the description will be oriented to the method of tuning brightness of the discharge tube. For example, for reducing the current IL, it is possible to shorten the conducting period of the switching elements Q 1  and Q 2 . 
     According to the present invention, the reference voltage is controlled so that the time taken in making the voltage of the capacitor C 1  or C 2  lower than the reference voltage Vref (VHL) is shortened. In FIG. 8, on the driving circuit on the low side, the comparator  8  is provided for comparing the voltage of the capacitor C 2  with the reference voltage Vref  2  and feeding a signal to the NAND circuit  7 . By making the reference voltage Vref 2  (VHL) of the comparator  8  higher than VHL at a normal lighting time by a dimming signal given on any timing, the conducting period of Q 2  may be reduced. The change of the reference voltage Vref 2  on the low side through this method makes it possible to tune light. 
     The resonant load circuit included in the lighting circuit is a current resonant type provided with the resonant inductor Lr and the capacitor Cr. If the discharge tube  16  is an electrodeless lamp, the discharge tube  16  operates to supply high-frequency current of several MHz to an exciting coil. It means that the inductor Lr used in the MHz-level high-frequency circuit becomes costly. The exciting coil takes a structure where a solenoid-type winding is wound around a magnetic substance. The coil equivalently corresponds to the inductor. The lighting circuit arranged to be usable as the exciting coil and the resonant inductor of the electrodeless lamp is illustrated in FIG.  10 . The discharge tube  16  is an electrodeless lamp having a winding wound around the magnetic substance. In FIG. 10, Lc denotes an equivalent inductor of the winding. The electrodeless lamp may be arranged to replace the winding of the exciting coil and the plasma generated inside of the discharge tube with a transformer shown in FIG. 11 as described in IEEE Transactions on Power Electronics Vol. 12, No.3, pp.507 to 516, 1997. In FIG. 11, the primary winding of the transformer corresponds to the winding of the exciting coil and the secondary winding corresponds to the plasma, in which the inductor is La and the equivalent resistor is Ra. 
     As described above, the equivalent circuit of the transformer-coupled electrodeless lamp is changed before and after the lighting circuit is operated. FIG. 12 shows the equivalent circuit before lighting. Since no plasma is generated inside of the discharge tube, the inductor Lc is a pure inductor composed of a coil wound around the magnetic substance. On the other hand, after lighting, since the generated plasma provides the equivalent inductance and resistance, the equivalent circuit is changed so that the circuit is connected in series to the inductor Ls and the resistor Rs as shown in FIG.  13 . The equivalent inductor La of the plasma makes the inductor Ls take a different value from the inductor shown in FIG. 12 Hence, the resonant frequency of the resonant load circuit before lighting is different from that after lighting. FIG. 14 shows a resonance curve of the resonant load circuit. As will be understood from FIG. 14, the equivalent inductance of the exciting coil after lighting is smaller than that before lighting. Hence, the resonant frequency fr 2  after lighting becomes higher than the resonant frequency fr 1  before lighting. 
     The lighting circuit arranged to be usable as the exciting coil and the resonant inductor of the electrode lamp provides the equivalent inductance of the exciting coil to be varied according to the lighting state of the lamp as mentioned above. Hence, as compared with the provision of the resonant inductor Lr, the great change of the resonant frequency of the resonant load circuit takes place. It means that the lighting circuit has to keep self-exciting in synchronous to the variation of the load. 
     FIG. 15 shows a lighting circuit arranged to be usable as the exciting coil and the resonant inductor of the electrodeless lamp. In FIG. 15, between the contact O and the N point are series-connected voltage drop means C 1 , C 2 , a resonant capacitor Cr, and an exciting coil and resonant inductor Lc of a discharge tube  16 . The switching elements Q 1  and Q 2  have their gate circuits arranged similarly with the foregoing circuit shown in FIG.  3  and their equivalent circuits arranged similarly with that shown in FIG.  5 . Next, the self-exciting drive will be described in synchronous to the variation of the load. FIG. 16 shows a phase difference φg between the resonant current and the gate voltage in the case of changing the resonant frequency of the resonant load circuit. As will be understood from FIG. 16, as the resonant frequency is higher, the phase difference φg is made smaller. This is because if the series circuit composed of Ciss, Rd 1  and L 1  has a capacitive impedance Zg, the increase of the frequency makes Zg come closer to an inductive characteristic. This makes the current flowing through the series circuit delayed and the gate voltage for the voltage of Ciss delayed. Hence, the phase difference φg of the gate voltage for the resonant load current is made smaller. For the change of the resonant frequency caused by the load variation, the gate circuit operates to automatically adjust the phase difference between the resonant current and the gate voltage and keep the self-exciting drive. That is, provision of the phase shift means in the gate circuit makes it possible to: 
     1) optionally adjust the on and off timing of Q 1  and Q 2 , thereby being able to change the operating frequency, and 
     2) follow the resonant condition if it is changed by the variation of the load. 
     In this system, when both of the gate voltages of the switching elements Q 1  and Q 2  come closer to the threshold voltage values of Q 1  and Q 2 , respectively, the shortcircuit may take place between these elements. Or, when Q 1  and Q 2  are turned on, if the gate voltage is turned on before the voltage between the drains and the sources of Q 1  and Q 2  are completely lowered to zero potential, Q 1  and Q 2  may be heated. FIG. 17 illustrates the latter phenomenon by using the waveforms of a voltage Vds between the drain and the source, a gate voltage Vg and a resonant current Il of the high-side gate circuit. As will be understood from FIG. 17, if the gate voltage Vg may take a waveform shown by a broken line, the element is heated as stated above. On the other hand, if the gate voltage Vg is changed to have a waveform indicated by a real line by delaying the turn-on of the gate voltage Vg, it is possible to prevent the elements from being heated. When Q 1  and Q 2  are turned on, by providing the waveform of the gate voltage with the delay time, it is possible to suppress the shortcircuit between the switching elements and heating. 
     FIG. 18 shows a gate circuit on the high side in which a gate voltage is provided with a delay period. The gate circuit on the low side has the same arrangement as the gate circuit on the high side. Hence, the figure of the high-side gate circuit is left out. In FIG. 18, the voltage drop means is a capacitor and the phase shift means is a series connection of an inductor and a resistor. The arrangement so far is the same as that shown in FIG.  15 . However, a capacitor Cd 1  is provided between the gate terminal and the drain terminal of Q 1 . In turn, the operation will be described. 
     When Q 2  is turned off, the resonant current is circulated in a diode QD 1  of Q 1  for reducing the voltage Vds between the drain and the source of Q 1 . In the case of flowing current to be charged into the capacitance between the gate and the source, the gate current bypasses the capacitor Cd 1 , which serves to suppress the rise of the gate voltage so that the delay period is provided. 
     As shown in FIG. 18, FIG. 19 shows the gate circuit on the high side in which the gate voltage is provided with the delay period. In FIG. 19, the voltage drop means and the phase shift means are the same as those shown in FIG. 15 but the diode Dg 1  is connected in parallel to the resistor Rd 1 . Dg 1  is directed so that the anode terminal is connected to the gate terminal of Q 1  and the cathode terminal is connected to a contact between the resistor Rd 1  and L 1 . The diode Dg 1  may be connected in parallel to L 1 . In this case, Dg 1  is directed so that the anode terminal is connected to a contact between the resistor Rd 1  and L 1  and the cathode terminal is connected to a contact between L 1  and C 1 . In FIG. 19, if the capacitance between the gate and the source is charged, the gate current is flown through the phase shift means L 1  and Rd 1  in sequence. On the other hand, if the charges in the capacitance between the gate and the source are discharged, the current is flown through the diode Dg 1  and L 1  in sequence. Thus, by changing the passage of the current discharged from the capacitance between the gate and the source, that is, by switching the impedance of the gate circuit, the delay may be provided when reversing the polarity of the gate voltage. 
     The lighting circuit shown in FIG. 20 provides a Zener diode ZD 2  between the gate terminal of Q 1  and one end of the capacitor C 1  and a Zener diode ZD 3  between the gate diode of Q 2  and one end of the capacitor C 2  and secures a dead time interval when the vertically located switching elements are turned on and off. For this type of gate circuit, the phase difference between the resonant current and the gate voltage is fixed. Hence, it is desirable to provide the resonant load circuit with the resonant inductor Lr so that the resonant frequency is made constant for the load variation. In FIG. 20, the voltage to be applied between the gate and the source of Q 1  is made to be a difference voltage between the voltage Vc 1  of the capacitor C 1  and the Zener voltage of ZD 2 . Likewise, the voltage to be applied between the gate and the source of Q 2  is made to be a difference voltage between the voltage Vc 2  and the Zener voltage of ZD 3 . Hence, when any one of the switching elements Q 1  and Q 2  is on, it is necessary to lower the gate voltage of the other switching element that is off by the Zener voltage of the Zener diode connected to the gate terminal of the switching element, for inserting a dead time between when Q 1  is on and when Q 2  is off. 
     The aforementioned embodiment is arranged to use the capacitors C 1  and C 2  for the voltage drop means Z 1  and Z 2 . In place, the voltage drop means Z 1  or Z 2  may be an inductor, a resistor or a combination of them. FIG. 21 shows a lighting circuit arranged to use resistors R 7  and R 8  for the voltage drop means Z 1  and Z 2 . In the case of connecting the resonant load circuit in series to the resistor, the maximum of the current flowing through the circuit is made smaller. Hence, it is desirable to set the resistance to a small value. The phase shift means L 1  or Rd 1  has the same arrangement as that shown in FIG. 15 but an input capacitance Ciss of Q 1  and a synthetic impedance of L 1  and Rd 1  are different from those of FIG.  15 . 
     In turn, the description will be oriented to the operation of the circuit shown in FIG. 21 with reference to FIG.  22 . FIG. 22 shows a waveform of each part of the gate circuit on the high side in the embodiment shown in FIG.  21 . In FIG. 22, the current IL flowing through the resonant load circuit causes the resistor R 7  to generate the voltage Vr 7  that is opposite in phase to IL with the O point as a reference level. If the input capacitance Ciss of Q 1  and the synthetic impedance of L 1  and Rd 1  are made inductive, the current IG 1  flowing through the phase shift means L 1  and Rd 1  is made to have a waveform of a delay phase rather than Vr 7 . The voltage to be applied between the gate and the source of q 1  is made to have a waveform whose phase is delayed by π/2 [rad] rather than Ig 1 . On the other hand, for the gate circuit on the low side, the voltage of the resistor R 8  is in phase with the resonant current IL. Hence, the low-side gate circuit operates in a reverse manner to the high-side gate circuit. 
     FIG. 23 is a lighting circuit arranged to use inductors L 3  and L 4  as voltage drop means Z 1  and Z 2 . In a case of connecting the inductors in series to the resonant load circuit, the resonant frequency of the load circuit is determined by a synthetic inductance containing the voltage drop means L 3  and L 4 . The phase shift means L 1  and Rd 1  are likewise to those shown in FIG.  15 . However, the synthetic impedance of Ciss, L 1  and Rd 1  of Q 1  is different from that shown in FIG.  15 . 
     In turn, the operation of the circuit shown in FIG. 23 will be described with reference to FIG.  24 . FIG. 24 represents the waveform of each part of the gate circuit on the high side included in the embodiment shonw in FIG.  23 . In FIG. 24, when the current IL is flown in the resonant load circuit, the voltage VL 3  of the inductor L 3  generated with an O point as a reference level is made to have a waveform of a lagged phase rather than IL. The current Ig 1  flowing the phase shift means L 1  and Rd 1  is made to have a waveform of a lagged phase rather than VL 3  if a synthetic impedance of the gate circuit composed of Ciss, Le and Rd 1  of Q 1  is inductive. The voltage to be applied between the gate and the source of Q 1  is made to have a waveform of a lagged phase rather than Ig 1  by π/2 [rad]. On the other hand, in the gate circuit on the low side, the voltage of the resistor R 8  has the same phase as the resonant current Il. Hence, the gate circuit on the low side performs in a reverse manner to the gate circuit on the high side. 
     FIG. 25 shows the embodiment arranged to apply the present invention to the conventional lighting circuit disclosed in JP-A-8-45685. In FIG. 25, a pair of switching elements Q 1  and Q 2  connected in series are connected between a positive and a negative electrodes of the voltage source  15 . A resonant inductor Lr, a resonant capacitor Cr, and a winding T 3  are connected in series between a contact O of those switching elements and a contact N between Q 2  and the negative electrode of the voltage source  15 . 
     A discharge tube  16  is provided as load in parallel to Cr and T 3 . The winding T 1  or T 2  is connected through the phase shift means Z 3  or Z 4  between the gate and the source of Q 1  or Q 2 . The Zener diodes coupled in series and in opposite to each other are provided in parallel to Q 1 . The winding T 1  has an opposite polarity to the winding T 2 . The windings T 1  and T 2  are magnetically coupled to a winding T 3 . The winding T 3  operates to sense the current flowing through the resonant load circuit and feed back the current to the windings T 1  and T 2  for doing the switching operation of Q 1  and Q 2 . The phase shift means Z 3  or Z 4  connected to the gate terminal of Q 1  or Q 2  provides impedance of a capacitor, an inductor, a resistor or a combination of them. By selecting the magnitude of the impedance, the on and off timing of Q 1  and Q 2  is optionally adjusted for changing the operating frequency. For providing a dead time between an on and an off of the vertically located switching elements, it is desirable to arrange the foregoing gate circuit as shown in FIG. 18, FIG. 19 or FIG.  20 . 
     FIG. 26 shows an embodiment in which the present invention is applied to the lighting circuit arranged to be usable as the exciting coil and the resonant inductor of the electrodeless lamp. In FIG. 26, a resonant capacitor Cr and an exciting coil and resonant inductor Lc of the discharge tube  16  are connected in series between the contacts O and N. A capacitor Ct and a winding T 3  connected in series are provided between both ends of Cr. The gate circuit of Q 1  or Q 2  has the same arrangement as shown in FIG.  25 . Hence, the description about the gate circuit is left out. In the resonant load circuit shown in FIG. 26, as mentioned above, the equivalent inductance of the exciting coil is changed according to the lighting state of the lamp. Hence, as compared with the provision of the resonant inductor Lr, the resonant frequency of the resonant load circuit is greatly changed. That is, if the load variation varies the resonant condition, the phase shift means Z 3  or Z 4  connected to the gate terminal serves to change the impedance for continuing the self-exciting drive. 
     FIG. 27 shows an embodiment in which the lighting circuit arranged to be usable as the exciting coil and the resonant inductor of the electrodeless lamp provides an N-channel switching element on the high side and a P-channel switching element on the low side. Between the contacts O and N are series-connected voltage drop means Z 1 , a resonant capacitor Cr, and an exciting coil and resonant inductor Lc of the discharge tube  16 . Phase shift means Z 3  is connected between one end of the voltage drop means Z 1  and the gate terminal of Q 1 . Phase shift means Z 4  is connected between one end of the voltage drop means Z 1  and the gate terminal of Q 2 . The voltage of Z 1  is applied to the gate terminals of Q 1  and Q 2  through Z 3  and Z 4 . In the case of applying an excessive voltage between the gate and the source of Q 1  or Q 2 , for preventing breakage of the switching element, it is possible to provide a Zener diode connected in series and in opposite to Q 1  or Q 2 . As mentioned above, by providing the low side with the P-channel switching element and making the lighting circuit complementary type, the voltage drop means connected to the resonant load circuit may be commonly used as the gate circuits of the vertical switching elements. As compared with the low side composed of the N-channel switching element, the number of parts is reduced in the low side of the P-channel switching element, which is effective in lowering the cost. In the case of providing two voltage drop means Z 1  and Z 2  in the resonant load circuit, it is considered that the variation of the parts may keep the vertical switching operation out of balance. By reducing the number of the voltage drop means into one, it is possible to solve this problem. The resonant load circuit shown in FIG. 27 is arranged to be usable as both the exciting coil and the resonant inductor of the electrodeless lamp. In place, an additional resonant inductor may be provided to the resonant load circuit. 
     The foregoing embodiment has concerned with the lighting circuit for supplying AC current to a current resonant type load circuit provided with the resonant inductor Lr and the capacitor Cr through the effect of the alternate switching operation of the pair of switching elements Q 1  and Q 2  connected in series. On the other hand, the lighting circuit for supplying an electric power to the resonant load circuit through the effect of one switching element is illustrated in FIG.  28 . The lighting circuit shown in FIG. 28 is arranged to be usable as both the exciting coil and the resonant inductor of the electrodeless lamp. Between a positive electrode and a negative one of a voltage source  15  are provided an inductor Lr and a capacitor Cp connected in series. The switching Q 1  is connected between both ends of the capacitor Cp. Letting O be a contact between Lr and Cp and N be a contact between Q 2  and the negative electrode of the voltage source  15 , the voltage drop means Z 1 , the resonant capacitor Cr, and the exciting coil and resonant inductor Lc of the discharge tube  16  are connected in series between the O and the N contacts. In FIG. 28, phase shift means Z 3  is provided between a gate terminal of the switching element Q 1  and the voltage drop means Z 1 . This phase shift means has a role of delaying or advancing a phase when applying the voltage of the voltage drop means between the gate and the source of Q 1 , for optionally adjusting the on and off timing of Q 1 . For the purpose of preventing breakage of the switching element caused by the application of an excessive voltage between the gate and the source of Q 1 , it is possible to provide a Zener diode connected in series and in opposite to Q 1 . In FIG. 28, the voltage drop means Z 1  and the phase shift means Z 3  may be a capacitor, an inductor, a resistor or a combination of them, for example. In a case that the resonant load circuit shown in FIG. 28 uses the resonant inductor, the voltage drop means Z 1 , the resonant inductor Lr, and the resonant capacitor Cr are connected in series between the O and the N points. The discharge tube  16  is provided in parallel to Cr. 
     FIG. 28 illustrates the lighting circuit arranged to use a single N-channel switching element, while FIG. 29 shows a lighting circuit arranged to use a P-channel switching element. In FIG. 29, a capacitor Cp and a switching element Q 1  connected in series are provided between a positive electrode and a negative one of a voltage source  15 . Between both ends of Cp are series-connected the voltage drop means Z 1 , a resonant capacitor Cr, and an exciting coil and resonant inductor Lc of the discharge tube  16 . The phase shift means Z 3  is provided between the gate terminal of the switching element Q 1  and the voltage drop means Z 1 . Further, a Zener diode may be provided in series and in opposite between the gate and the source of Q 1 . In a case that the resonant load circuit shown in FIG. 29 uses the resonant inductor, the voltage drop means Z 1 , the resonant inductor Lr, and the resonant capacitor Cr are connected in series between both ends of Cp. The discharge tube  16  is provided in parallel t Cr. 
     The aforementioned embodiment has been arranged to connect the voltage drop means in series to the resonant load circuit and drive the switching element in response to the voltage caused by the resonant current. On the other hand, the following description will be oriented to an embodiment in which the switching elements and the built-in diodes are driven by the voltage caused by the resonant current only when they are on. 
     In the embodiment shown in FIG. 30, an AC power source AC supplies AC current to a rectifying circuit composed of a diode bridge DB through an AC filter composed of inductors Lf and Cf. The rectified current is sent to a voltage source  15 . The voltage source  15  operates to generate a DC voltage from the rectified current. A drain terminal of Q 1  is connected to a positive electrode of the voltage source. A capacitor C 1  is connected as voltage drop means between a source terminal of Q 1  and a drain terminal of Q 2 . The contact between C 1  and Q 2  is denoted by O. The capacitor C 1  is connected in parallel to a resistor R 1 . Further, a capacitor C 2  is connected between a source terminal of Q 2  and a negative electrode of the voltage source  15 . The contact between Q 2  and the negative electrode of the voltage source  15  is denoted by N. Like C 1 , the capacitor C 2  is connected in parallel to a resistor R 2 . Between the contacts O and N are series-connected a resonant inductor and a resonant capacitor Cr. Cr provides a discharge tube  16  in parallel thereto. If the discharge tube is an electrodeless lamp, Lr shown in FIG. 30 may be usable as the exciting coil and the resonant inductor. 
     A capacitor C 3  is connected in parallel between the drain terminal and the source terminal of Q 1 . A gate resistor R 5  and a capacitor  13  are connected in series between the gate terminal and the output O. Like Q 1 , a capacitor C 4  is also connected in parallel between the drain terminal and the source terminal of Q 2 . A gate resistor R 6  and a capacitor  14  are connected in series between the gate terminal and the contact N. The capacitor  14  is connected in parallel to a Zener diode ZD 1 . Further, a resistor Rs 3  is connected to a contact between the inductors Lf and Cf connected in series to the cathode of the diode ZD 1 . Herein, the voltage of the capacitor  14  serves to drive Q 2  so that the Zener voltage of the diode ZD 1  may be kept irrespective of the magnitude of the AC voltage. Further, the voltage of the capacitor  13  serves to drive Q 1 . The capacitor  13  is discharged by the voltage of the capacitor C 14 . With Q 2  being turned on, the voltage of the capacitor C 14  with the N point as a reference level is supplied to the capacitor  13  through the diode D 1 . 
     In turn, the description will be oriented to the operation of this circuit with reference to FIG.  31 . FIG. 31 shows a waveform of each part included in the embodiment shown in FIG.  30 . The current IL of a resonant circuit is defined assuming that the flowing direction of the current IL of the resonant circuit from the O point shown in FIG. 30 is positive. In this definition, during one period of the current IL, four operation modes are provided about Q 1 , A 2 , QD 1  and QD 2 . These periods of the modes are indicated as t 1  to t 4 . Hereafter, each operation mode will be described. 
     Mode 1 (t 1  period): When Q 1  is turned on, the voltage source  15  operates to flow the current IL through a passage of Q 1 , C 1 , Lr and Cr. The current IL is charged in Cr and also is partially branched into the discharge tube  16 . Further, the capacitor C 1  is charged by the current IL. The voltage applied between the gate and the source of Q 1  at the mode 1 corresponds to a difference voltage between the voltage of the capacitor  13  and Vc 1 . With increase of Vc 1 , the gate voltage of Q 1  is decreased. When the gate voltage is equal to or lower than a threshold voltage of MOSFET, Q 1  is turned off. 
     During this period, the capacitor C 2  is discharged through the effect of the resistor R 2 . The voltage Vc 2  of C 2  is gradually decreased as shown in FIG.  31 . 
     Q 1  has a capacitor C 3  located in parallel thereto and Q 2  also has a capacitor C 4  located in parallel thereto. When Q 1  is turned off, as shown in FIG. 9, the current IL/ 2  is caused to flow through C 3  and a voltage rise dV/dt between the drain and the source of Q 1  is limited by IL/ 2 C 3 . At a time, the flow of the current IL/ 2  is caused to discharge C 4  and dV/dt at the voltage drop of Q 2  is similarly limited by Il/ 2 C 4 . At the switching time, dV/dt may cause conduction noises and radiation noises. Like this embodiment, however, by doing the soft switching for suppressing dV/dt, it is possible to lighten this problem. Further, a current waveform of Q 1  appearing when the t 1  period is terminated is denoted by a circle mark. At this time point, the voltage of Q 1  is substantially zero. It means that no switching loss exists which loss might be caused if the current of Q 1  is overlapped with the voltage thereof. The soft switching is effective in reducing the switching loss. 
     At the operation, the current charged in C 3  is charged in C 1  and the gate voltage of Q 1  is further decreased, so that Q 1  may be stably turned off. On the other hand, the current to be discharged from the capacitor C 4  is conversely charged to C 2  so that Vc 2  is further reduced. 
     Mode 2 (t 2  period): When Q 1  is turned off, the current IL has a value of a positive polarity. This current is kept flowing through a passage of Lr, Cr, C 2  and QD 2 . Part of the current IL is branched into the discharge tube  16 . 
     The current IL serves to reversely charge C 2  so that Vc 2  is reduced. Like Q 1 , the voltage to be applied between the gate and the source of Q 2  corresponds to a difference voltage between the voltage of the capacitor  14  and Vc 2 . With decrease of Vc 2 , the gate voltage of Q 2  is increased. When the gate voltage is greater than or equal to a threshold voltage of MOSFE, Q 2  is turned on. During the mode 2 period, the current polarity is opposite as viewed from Q 2 . As shown in FIG. 31, if the gate voltage is charged, the current is kept flowing through QD 2  as long as the current polarity is constant. The period to the change of the polarity of the current IL to a negative one corresponds to the mode 2, during which the reverse charge of C 2  is continued so that Vc 2  may be reduced. 
     During the mode 2, the capacitor C 1  is caused to be discharged by the resistor R 1  so that Vc 1  is progressively reduced. 
     Mode 3 (t 3  period): When the polarity of the current IL is changed from a positive value to a negative value, the current IL is kept flowing through Q 2  whose gate voltage is charged at the mode 2. That is, the current IL is flown as discharge current through a passage of Q 2 , C 2 , Cr and Lr. C 2  is charged by the current IL. Vc 2  is increased by IL. When the gate voltage is equal to or lower than the threshold voltage of MOSFET, Q 2  is turned off. At the mode 3, like the mode 1, while the polarity of the current IL is negative, Q 2  is turned off. 
     During the mode 3, the capacitor C 1  is discharged by the resistor R 1 , so that Vc 1  is progressively reduced. 
     When Q 2  is turned off, like when Q 1  is turned off, the capacitors C 3  and C 4  located in parallel to Q 1  and Q 2  cause the current IL/ 2  to flow through C 4 . Then, the voltage rise dV/dt between the drain and the source of Q 2  is limited by IL/ 2 C 4 . At a time, the flow of the current IL/ 2  causes C 3  to be discharged. Like Q 2 , dV/dt at the voltage drop of Q 1  is limited by IL/ 2 C 3 . The current waveform of Q 2  appearing when the t 3  period is terminated is denoted by a circle mark. At this time point, the voltage of Q 2  is made substantially zero. It means that no switching loss exists which loss might be caused if the current of Q 2  is overlapped with the voltage thereof. 
     In the foregoing operation, the current charged in C 4  serves to charge C 2  and the gate voltage of Q 2  is further reduced. Hence, Q 2  may be stably turned off. The current charged in C 3  is reversely charged in C 1 , so that Vc 1  is decreased. 
     Mode 4 (t 4  period): When Q 2  is turned off, the current IL has a value of a negative polarity. The electromagnetic energy stored in Lr serves to keep the current IL flowing through a passage of Lr, C 1 , QD 1 , the voltage source  15 , Cr and Lr. In addition, part of the current IL is branched to the discharge tube  16 . 
     The current IL has an effect on reversely charging C 1 , thereby reducing Vc 1  and increasing the gate voltage of Q 1 . When the gate voltage is higher than or equal to the threshold value of MOSFET, Q 1  is turned on. During the mode 4 period, the current polarity is opposite as viewed from Q 1 . If the gate voltage is charged, the current is kept flowing through QD 1  as long as the polarity of the current is constant. The period to the change of the polarity of the current IL to a positive polarity corresponds to the mode 4. During this period, the reverse charge of C 1  is continued so that Vc 1  is reduced. Further, during the period of mode 4, the capacitor C 2  is discharged by the resistor R 2  so that Vc 2  is gradually decreased. 
     The resistors R 1  and R 2  connected in parallel to the capacitors C 1  and C 2  have an effect on applying a bias voltage onto a voltage of the capacitor. This has a role of lowering the gate voltage of MOSFET, thereby providing a dead time interval between an on state and an off state of the vertically located switches. Further, by decreasing the voltage of the capacitor while the switch is turned off, it is possible to adjust the on periods of the vertically located switches. 
     As set forth above, during one period of the current IL, the operations from the mode 1 to the mode 4 are executed. Later, these operations are repeated. 
     The present system is arranged to turn off Q 1  and Q 2  according to the capacitor voltages Vc 1  and Vc 2  and has a shortcoming of lowering a current performance as the gate voltages of Q 1  and Q 2  are decreased and come closer to zero. This means the increase of an on resistance. In particular, in a case that the driving frequency is about several tens kHz, it is desirable to apply a great gate voltage enough to inhibit the increase of the on resistances of Q 1  and Q 2  if the switching loss of Q 1  and Q 2  is greater than a constant loss. Hence, an embodiment for solving this kind of problem is shown in FIG.  32 . 
     The arrangement of a resonant load circuit included in the embodiment shown in FIG. 32 is likewise to that included in the embodiment shown in FIG.  30 . Hence, the description thereabout is left out. As will understood from FIG. 32, the gate terminal of Q 1  is connected with the drain terminal of MOSFET S 3  and a resistor R 7  is connected between the source terminal of Q 1  and the gate terminal of Q 2 . The source terminal of S 3  is connected to a contact O. The arrangement of the driving circuit of Q 2  is likewise to that of Q 2 . The gate terminal of Q 2  is connected with the drain terminal of MOSFET S 4  and a resistor R 8  is connected between the source terminal of Q 2  and the gate terminal of S 4 . The source terminal of S 4  is connected with a contact N. 
     In turn, the operation of the embodiment shown in FIG. 32 will be described with reference to FIG.  33 . FIG. 33 illustrates a waveform of each part of FIG.  32 . 
     At first, when Q 1  is turned on, the voltage source  15  operates to flow current IL through a passage of Q 1 , C 1 , Lr and Cr. The current IL is charged in the capacitor C 1 . When Vc 1  exceeds a threshold voltage of MOSFET S 3 , S 3  is turned on. This makes the gate voltage of Q 1  discharged through S 3  and C 1  and the voltage Vc 1  applied between the source and the gate in a reversely biased manner. This results in preventing Q 1  from being minutely turned on again by the adverse factors such as noises, thereby being able to stably turn off Q 1 . 
     Next, when Q 1  is turned off, the same operation as that of the mode 2 described with reference to FIGS. 30 and  31  is started. The description thereabout is likewise to the foregoing description. Hence, it is left out. 
     When the polarity of the current IL is changed from a positive polarity to a negative one, the current IL is caused to flow through Q 2 . That is, the current IL is flown as discharge current of Cr through a passage of Q 2 , C 2 , Cr and Lr and C 2  is charged by IL. IL serves to increase Vc 2 . When Vc 2  exceeds a threshold voltage of MOSFET S 4 , S 4  is turned on. This makes the gate voltage of Q 2  discharged through S 4  and C 2  and the voltage Vc 2  applied between the source and the gate of Q 2  in a reversely biased manner, thereby turning off Q 2 . This period corresponds to the mode 3. 
     When Q 2  is turned off, the same operation as that of the mode described with reference to FIGS. 30 and 31 is started. The description of the operation of this period is likewise to the foregoing one. Hence, it is left out. 
     The foregoing operation is executed during one period of the current IL. After that, this operation is repeated. 
     In turn, a voltage resonance type lighting circuit arranged to use one switching element is illustrated in FIG.  34 . As shown in FIG. 34, a resonant capacitor Cp is connected between a positive electrode of the voltage source  15  and a drain terminal of the switching element Q 2 . Further, between both ends of the capacitor Cp are series-connected a resonant inductor Lr and a resonant capacitor Cr. Cr provides a discharge tube  16  as a load in parallel thereto. A resonant load circuit is not limited to the arrangement shown in FIG.  34 . Lr may be usable as the exciting coil and the resonant inductor of the electrodeless lamp. A capacitor C 2  is connected between a source terminal of Q 2  and a negative electrode of the voltage source  15 . A resistor R 2  is connected in parallel to C 2 . Further, the resistor R 2  is connected in parallel to C 2 . Further, the capacitor C 4  is connected between the drain and the source terminals of Q 2 . Letting the negative electrode of the voltage source  15  be an N point, a gate resistor R 6  and a capacitor  14  are connected in series between the gate terminal and the N point, and a Zener diode ZD 1  is connected in parallel to the capacitor  14 . Further, a resistor Rs 3  is connected between a cathode of the diode ZD 1  and a contact between the inductors Lf and Cf connected in series to each other. 
     In turn, the description will be oriented to the operation shown in FIG.  34 . At first, when Q 2  is turned on, the voltage source  15  is caused to flow the current IL through Lr, Cr, Q 2 , and C 2  so that the capacitor C 2  is charged. During this period, the voltage to be applied between the gate and the source corresponds to a difference voltage between the voltage of the capacitor  14  and Vc 2 . With increase of Vc 2 , the gate voltage of Q 2  is decreased. When the gate voltage is lower than or equal to the threshold voltage of MOSFET, Q 2  is turned off. During this period, the voltage of the capacitor Cp corresponds to the voltage of the voltage source  15 . 
     When Q 2  is turned off, C 4  located in parallel to Q 2  serves to branch the current IL into Cp and C 4 . Assuming that the current of C 4  is denoted by Ic 4 , a voltage rise dV/dt between the drain and the source of Q 2  is limited by Ic 4 ×C 4 . Further, the current charged in C 4  is charged into C 2  and the gate voltage of Q 2  is further decreased. This makes it possible to stably turn off Q 2 . In the case of removing the capacitor C 4 , with increase of a drain voltage of Q 2 , the voltage of the capacitor C 2  is increased and the voltage of the capacitor C 2  is increased as well. On the other hand, the gate voltage of Q 2  is decreased. Hence, Q 2  is never turned on again after Q 2  is turned off. At a time point when Q 2  is turned off, the current IL is caused to flow through a resonant passage of Lr, Cr and Cp, which serves to progressively decrease the voltage of the capacitor Cp. Further, the voltage between the drain and the source of Q 2  is substantially zero. It means that no switching loss exists which loss might be caused if the current of Q 2  is overlapped with the voltage of Q 2 . 
     In turn, when Q 2  is turned off, the current IL is kept flowing through a passage of Lr, Cr and Cp. During this period, the electromagnetic energy accumulated in Lr causes the current IL to flow through the passage. Hence, the current is kept flowing through Cp and reversely charged in Cp as long as the current polarity is constant. On the other hand, the capacitor C 2  is discharged by the resistor R 2  and the voltage is progressively decreased. This operation is continued until the polarity of the current IL is changed into a negative one. 
     The change of the polarity of the current IL causes the current IL to flow as discharge current through a passage of Cp, Cr and Lr. The flow of the current Il is continued until the voltage of CP reaches the voltage of the voltage source  15 . On the other hand, the voltage between the drain and the source of Q 2  is gradually decreased with increase of the voltage of the capacitor Cp. 
     When the voltage of the capacitor Cp reaches the voltage of the voltage source  15 , the current IL is caused to flow through C 4  located in parallel to Q 2 . The current of C 4  is reversely charged in C 2  and the gate voltage of Q 2  is increased. In succession, the electromagnetic energy stored in Lr causes the current IL to continuously flow through a passage of Lr, the voltage source  15 , C 2 , QD 2  and Cr. C 2  is reversely charged as long as the current polarity is constant. When the gate voltage exceeds the threshold voltage of MOSFET, Q 2  is turned on. 
     During one period of the current IL, the foregoing operation is executed. Later, this operation is repeated. 
     In the embodiment shown in FIG. 34, the driving circuit of Q 2  is arranged as shown in FIG.  32 . This arrangement makes it possible to solve the problem of progressively decreasing the gate voltage of Q 2  with increase of the capacitor voltage Vc 2 , thereby reducing the current performance. 
     In the case of building the lighting circuit in a connector of an incandescent lamp, the heat of the discharge tube raises the temperature inside of the connector up to about 100° C. Hence, the lighting circuit has to keep the constant operation even under the high temperature environment. The phase shift means connected to the gate terminal of the switching element may be used as impedance for compensating for a temperature by changing an impedance value at a high temperature. The present invention enables to stabilize the operation even at a high temperature. It means that the present invention may be also suitable to a bulb type fluorescent lamp having the lighting circuit built in the connector of the light. 
     The lighting device for illumination according to the present invention makes it possible to guarantee a stable resonant operation in synchronous to a resonant frequency of a load and a current even when the resonant conditions are changed by varying a resonant load composed of a discharge tube, a resonant inductor and a resonant capacitor. Further, since the lighting circuit may be composed of inexpensive parts, the present invention is economical.