Patent Publication Number: US-8977542-B2

Title: Audio encoder and decoder and methods for encoding and decoding an audio signal

Description:
TECHNICAL FIELD 
     The present invention relates to the field of audio signal encoding and decoding. 
     BACKGROUND 
     A mobile communications system presents a challenging environment for voice transmission services. A voice call can take place virtually anywhere, and the surrounding background noises and acoustic conditions will have an impact on the quality and intelligibility of the transmitted speech. At the same time, there is strong motivation for limiting the transmission resources consumed by each communication device. Mobile communications services therefore employ compression technologies in order to reduce the transmission bandwidth consumed by the voice signals. Lower bandwidth consumption yields lower power consumption in both the mobile device and the base station. This translates to energy and cost saving for the mobile operator, while the end user will experience prolonged battery life and increased talk-time. Furthermore, with less consumed bandwidth per user, a mobile network can service a larger number of users at the same time. 
     Today, the dominating compression technology for mobile voice services is Code Excited Linear Prediction (CELP), described for example in “Code-Excited Linear Prediction (CELP) high-quality speech at very low bit rates”, M. R. Schroeder and B. Atal, IEEE ICASSP 1985. 
     CELP is an encoding method operating according to an analysis-by-synthesis procedure. In CELP for voice coding, linear prediction analysis is used in order to determine, based on an audio signal to be encoded, a slowly varying linear prediction (LP) filter A(z) representing the human vocal tract. The audio signal is divided into signal segments, and a signal segment is filtered using the determined A(z), the filtering resulting in a filtered signal segment, often referred to as the LP residual. A target signal x(n) is then formed, typically by filtering the LP residual through a weighted synthesis filter W(z)/Â(z) to form a target signal x(n) in the weighted domain. The target signal x(n) is used as a reference signal for an analysis-by-synthesis procedure wherein an adaptive code book is searched for a sequence of past excitation samples which, when filtered through weighted synthesis filter, would give a good approximation of the target signal. A secondary target signal x 2 (n) is then derived by subtracting the selected adaptive code book signal from the filtered signal segment. The secondary target signal is in turn used as a reference signal for a further analysis-by-synthesis procedure, wherein a fixed code book is searched for a vector of pulses which, when filtered through the weighted synthesis filter, would give a good approximation of the secondary target signal. The adaptive code book is then updated with a linear combination of the selected adaptive code book vector and the selected fixed code book vector. 
     By use of CELP, a good speech quality at moderately low bandwidth is typically achieved, and the method is widely used in deployed codecs such as GSM-EFR, AMR and AMR-WB. However, for the very low bit rates, the limitations of the CELP coding technique begin to show. While the segments of voiced speech remain well represented, the more noise-like consonants such as fricatives start to sound worse. Degradation can also be perceived in the background noises. 
     As seen above, the CELP technique uses a pulse based excitation signal. For voiced signal segments, the filtered signal segment (target excitation signal) is concentrated around so called glottal pulses, occurring at regular intervals corresponding to the fundamental frequency of the speech segment. This structure can be well modeled with a vector of pulses. For a noise-like segment, on the other hand, the target excitation signal is less structured in the sense that the energy is more spread over the entire vector. Such an energy distribution is not well captured with a vector of pulses, and particularly not at low bitrates. When the bit rate is low, the pulses simply become too few to adequately capture the energy distribution of the noise-like signals, and the resulting synthesized speech will have a buzzing distortion, often referred to as the sparseness artefact of CELP codecs. 
     Hence, for the very low bit rates, which could for example be advantageous when the transmission channel conditions are poor, an alternative to the CELP is required in order to arrive at a well sounding synthesized signal. Several technologies have been developed in order to deal with the CELP sparseness artefact at low bitrates. WO99/12156 discloses a method of decoding an encoded signal, wherein an anti-sparseness filter is applied as a post-processing step in the decoding of the speech signal. Such anti-sparseness processing reduces the sparseness artefact, but the end result can still sound a bit unnatural. 
     Another method of mitigating the sparseness artefact which is well known in the art is often referred to as Noise Excited Linear Prediction (NELP). In NELP, signal segments are processed using a noise signal as the excitation signal. The noise excitation is only suitable for representation of noise-like sounds. Therefore, a system using NELP often uses a different excitation method, e.g. CELP, for the tonal or voiced segments. Thus, the NELP technology relies on a classification of the speech segment, using different encoding strategies for unvoiced and voiced parts of an audio signal. The difference between these coding strategies gives rise to switching artefacts upon switching between the voiced and unvoiced switching strategies. Furthermore, the noise excitation will typically not be able to successfully model the excitation of complex noise-like signals, and parts of the anti-sparseness artefacts will therefore typically remain. 
     As can be seen from the above, there is a need for an improved codec by which a high quality synthesized audio signal can be obtained even when the encoded signal is encoded for low bit rate transmission. 
     SUMMARY 
     An object of the present invention relates is to improve the quality of a synthesized audio signal when the encoded signal is transmitted at a low bit rate. 
     This object is addressed by an encoding method, a decoding method, an audio encoder, an audio decoder, and computer programs for encoding and decoding of an audio signal. 
     A method of encoding and decoding an audio signal is provided, wherein an adaptive spectral code book of an encoder, as well as of a decoder, is updated with frequency domain representations of encoded time domain signal segments. A received time domain signal segment is analysed by an encoder to yield a frequency domain representation, and an adaptive spectral code book in the encoder is searched for an ASCB vector which provides a first approximation of the obtained frequency domain representation. This ASCB vector is selected. A residual frequency representation is generated from the difference between the frequency domain representation and the selected ASCB vector. A fixed spectral code book in the encoder is then searched for an FSCB vector which provides an approximation of the residual frequency representation. This FSCB vector is also selected. A synthesized frequency representation may be generated from the two selected vectors. The encoder further generates a signal representation indicative of an index referring to the selected ASCB vector, and of an index referring to the selected FSCB vector. The gains of the linear combination can advantageously also be indicated in the signal representation. 
     A signal representation generated by an encoder as discussed above, can be decoded by identifying, using the ASCB index and FSCB index retrieved from the signal representation, an ASCB vector and an FSCB vector. In decoding of the signal representation, a linear combination of the identified ASCB vector and the identified FSCB vector provides a synthesized frequency domain representation of the time domain signal segment to be synthesized. A synthesized time domain signal is generated from the synthesized frequency domain representation. 
     By using a frequency domain representation of a time domain signal segment in the encoding of an audio signal, control of the spectral distribution of noise-like sounds can efficiently be obtained also at low bitrates, and the synthesis of such sounds can thereby be improved when the transmission channel between the encoder and decoder provides a low bitrate. Since the length of the time domain signal segments considered for encoding of speech signals is relatively short, the corresponding frequency domain representation will likely show large variations between time-adjacent frames. By providing an adaptive spectral code book which is frequently updated, it is ensured that a suitable approximation of the frequency domain representation can be found, despite the anticipated poor correlation between time-adjacent frequency domain representations of time domain signal segments. 
     In one embodiment, the frequency domain representation is obtained by performing a time-to-frequency domain transformation analysis of a time domain signal segment, thereby obtaining a segment spectrum. The frequency domain representation is obtained as at least a part of the segment spectrum. The time-to-frequency domain transform could for example be a Discrete Fourier Transform (DFT), where the obtained segment spectrum comprises a magnitude spectrum and a phase spectrum. The frequency domain representation could then correspond to the magnitude spectrum part of the segment spectrum. Another example of a time-to-frequency domain transform analysis is the Modified Discrete Cosine Transform analysis (MDCT), which generates a single real-valued MDCT spectrum. In this case, the frequency domain representation could correspond to the MDCT spectrum. Other analyses may alternatively be used. In another embodiment, the frequency domain representation is obtained by performing a linear prediction analysis of a time domain signal segment. 
     In one embodiment, the encoding/decoding method applied to a time domain signal segment is dependent on the phase sensitivity of the sound information carried by the segment. In this embodiment, an indication of whether a segment should be treated as phase insensitive or phase sensitive could be sent to the decoder, for example as part of the signal representation. For a segment which carries phase insensitive information, the generation of a synthesized time domain signal from the synthesized frequency domain representation could include a random component, which could advantageously be generated in the decoder. For example, when the frequency analysis performed in the encoder is a DFT, the phase spectrum could be randomly generated in the decoder; or when the frequency analysis is an LP analysis, a time domain excitation signal could be randomly generated in the decoder. For the encoding of a segment carrying phase sensitive information, a time domain based encoding method, such as CELP, would be used. Alternatively, a frequency domain based encoding method using an adaptive spectral code book could be used also for encoding of phase sensitive signal segments, where the signal representation includes more information for phase sensitive signal segments than for phase insensitive. For example, if some information is randomly generated in the decoder for phase insensitive segments, at least part of such information can, for phase sensitive segments, instead be parameterized by the encoder and conveyed to the decoder as part of the signal representation. 
     By using different encoding/decoding methods for different types of sounds, the bandwidth requirements for the transmission of the signal representation can be kept low, while allowing for the noise like sounds to be encoded by means of a frequency domain based encoding method using an adaptive spectral code book. 
     Randomly generated information, such as the phase of a segment spectrum or a time domain excitation signal, could in one embodiment be used for all signal segments, regardless of phase sensitivity. 
     When the frequency analysis is a DFT and a randomly generated phase spectrum is used in the decoding of a segment, the sign of the DC component of the random spectrum can for example be adjusted according to the sign of the DC component of the segment spectrum, thereby improving the stability of the energy evolution between adjacent segments. Hence, the sign of the DC component of the segment spectrum can be included in the signal representation. By using randomly generated phase information when synthesizing the segment spectrum, the amount of phase information that has to be transmitted from the encoder to the decoder can be greatly reduced or, in some embodiments, even eliminated. 
     The encoding method may, in one embodiment, include an estimate of the quality of the first approximation of the frequency domain representation. If such quality estimation indicates the quality to be insufficient, the encoder could enter a fast convergence mode, wherein the frequency domain representation is approximated by at least two FSCB vectors, instead of one FSCB vector and one ASCB vector. This can be useful in situations where the audio signal to be encoded changes rapidly, or immediately after the adaptive spectral code book has been initiated, since the ASCB vectors stored in the adaptive spectral code book may then be less suitable for approximating the frequency domain representation. The fast convergence mode can be signaled to the decoder, for example as part of the signal representation. The adaptive spectral code book of the encoder and of the decoder can advantageously be updated also in the fast convergence mode. 
     The updating of the adaptive spectral code book of the encoder and of the decoder can be conditional on a relevance indicator exceeding a relevance threshold, the relevance indicator providing a value of the relevance of a particular frequency domain representation for the encodability of future time domain signal segments. The global gain of a segment could for example be used as a relevance indicator. In the decoder, the value of the relevance indicator could in one implementation be determined by the decoder itself, or a value of the relevance indicator could be received from the encoder, for example as part of the signal representation. 
     Further aspects of the invention are set out in the following detailed description and in the accompanying claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic illustration of an audio codec system comprising an encoder and a decoder. 
         FIG. 2  is a flowchart illustrating a method of encoding an audio signal into a signal representation. 
         FIG. 3  is a flowchart illustrating a method of decoding a signal representation and synthesizing an audio signal. 
         FIG. 4  schematically illustrates an embodiment of an audio encoder. 
         FIG. 5  schematically illustrates an embodiment of an audio decoder. 
         FIG. 6  is a flowchart illustrating a feature of an embodiment of the encoding and decoding methods. 
         FIG. 7  schematically illustrates a feature of an embodiment of the codec. 
         FIG. 8  is a flowchart illustrating a feature of an embodiment of the encoding method. 
         FIG. 9  schematically illustrates a feature of an embodiment of the encoder. 
         FIG. 10  schematically illustrates a decoder feature corresponding to the encoder feature shown in  FIG. 9 . 
         FIG. 11  is a flowchart illustrating a feature of an embodiment of the encoding method, whereby the encoder can enter one of a phase sensitive of phase insensitive encoding modes. 
         FIG. 12  is a flowchart illustrating an embodiment of the encoding method of  FIG. 2 . 
         FIG. 13  is a flowchart illustrating an embodiment of the decoding method of  FIG. 3 . 
         FIG. 14  schematically illustrates an embodiment of an encoder. 
         FIG. 15  schematically illustrates an embodiment of a decoder. 
         FIG. 16  schematically illustrates an embodiment of an encoder. 
         FIG. 17  schematically illustrates an embodiment of a decoder. 
         FIG. 18  is an alternative illustration of an encoder or of a decoder. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  schematically illustrates a codec system  100  including a first user equipment  105   a  having an encoding  110 , as well as a second user equipment  105   b  having a decoder  112 . A user equipment  105   a/b  could, in some implementations, include both an encoder  110  and a decoder  112 . When generally referring to any user equipment, the reference numeral  105  will be used. 
     The encoder  110  is configured to receive an input audio signal  115  and to encode the input signal  115  into a compressed audio signal representation  120 . The decoder  112 , on the other hand, is configured to receive an audio signal representation  120 , and to decode the audio signal representation  120  into a synthesized audio signal  125 , which hence is a re-production of to the input audio signal  115 . The input audio signal  115  is typically divided into a sequence of input signal segments, either by the encoder  110  or by further equipment prior to the signal arriving at the encoder  110 , and the encoding/decoding performed by the encoder  110 /decoder  112  is typically performed on a segment-by-segment basis. Two consecutive signal segments may have a time overlap, so that some signal information is carried in both signal segments, or alternatively, two consecutive signal segments may represent two distinctly different, and typically adjacent, time periods. A signal segment could for example be a signal frame, a sequence of more than one signal frames, or part of a signal frame. 
     According to the invention, the effects of sparseness artefacts at low bitrates discussed above in relation to the CELP encoding technique can be avoided by using an encoding/decoding technique wherein an input audio signal is transformed, from the time domain, into the frequency domain, so that a signal spectrum is generated. By introducing the possibility of directly controlling the spectral energy distribution of a signal segment, the noise-like signal segments can be more accurately reproduced even at low bitrates. A signal segment which carries information which is aperiodic can be considered noise-like. Examples of such signal segments are signal segments carrying fricative sounds and noise-like background noises. 
     Transforming an input audio signal into the frequency domain as part of the encoding process is know from e.g. WO95/28699 and “ High Quality Coding of Wideband Audio Signals using Transform Coded Excitation  ( TCX )”, R. Lefebvre et al.,  ICASSP  1994, pp. I/193-I/196 vol. 1. The method disclosed in these publications, referred to as TCX and wherein an input audio signal is transformed into a signal spectrum in the frequency domain, was proposed as an alternative to CELP at high bitrates where CELP requires high processing power—the computation requirement of CELP increases exponentially with bitrate. 
     In the TCX encoding method of R. Lefebvre et al, a prediction of the signal spectrum is given by the previous signal spectrum, obtained from transforming the previous signal segment. A prediction residual is then obtained as the difference between the prediction of the signal spectrum and the signal spectrum itself. A spectral prediction residual code book is then searched for a residual vector which provides a good approximation of the prediction residual. 
     The TCX method has been developed for the encoding of signals which require a high bitrate and wherein a high correlation exists in the spectral energy distribution between adjacent signal segments. An example of such signals is music. For signal segments representing noise-like sounds such as fricatives, on the other hand, the spectral energy distribution of adjacent signal segments are generally less correlated when using segment lengths typical for voice encoding (where e.g. 5 ms is an often used duration of a voice encoding signal segment). A longer signal segment time duration is often not appropriate, since a longer time window will reduce the time resolution and possibly have a smearing effect on noise-like transient sounds. 
     According to the invention, control of the spectral distribution of noise-like sounds can, however, be obtained by using an encoding/decoding technique wherein a time domain signal segment originating from an audio signal is transformed into the frequency domain, so that a segment spectrum is generated, and wherein an adaptive spectral code book (ASCB) is used to search for a vector which can provide an approximation of the segment spectrum. The ASCB comprises a plurality of adaptive spectral code book vectors representing previously synthesized segment spectra, of which one, which will provide a first approximation of the segment spectrum, is selected. A residual spectrum, representing the difference between the segment spectrum and the first spectrum approximation, is then generated. A fixed spectral code book (FSCB) is then searched to identify and select a FSCB vector which can provide an approximation of the residual spectrum. The signal segment can then be synthesized by use of a linear combination of the selected ASCB vector and the selected FSCB vector. The ASCB is then updated by including a vector, representing the synthesized magnitude spectrum, in the set of spectral adaptive code book vectors. 
     By use of a time-vs-frequency domain transform in combination with an adaptive spectral code book for encoding an audio signal segment is achieved that an efficient encoding and decoding of audio signals can be obtained, wherein noise-like sounds are reproduced in a satisfying manner. Experimental studies show that, although adaptive code books in time domain are typically used to facilitate the encoding of strongly periodic signals, the encoding of noise-like signals, which are typically aperiodic, can be efficiently performed by use of an adaptive spectral code book. The time-vs-frequency domain transform facilitates for the accurate control of the spectral energy distribution of a signal segment, while the adaptive spectral code book ensures that a suitable approximation of the segment spectrum can be found, despite possible poor correlation between time-adjacent segment spectra of signal segments carrying the noise-like sounds. 
     An encoding method according to an embodiment of the invention is shown in  FIG. 2 . The method shown in  FIG. 2  will be referred as a transform based adaptive encoding method. At step  200 , a time domain (TD) signal segment  T   m  comprising N samples is received at an encoder  110 , where m indicates a segment number. In the following description of  FIGS. 2 and 3 , the encoding and decoding of a particular signal segment is described, and the segment number m will be omitted from the description. The TD signal segment  T  can for example be a segment of an audio signal  115 , or the TD signal segment can be a quantized and pre-processed segment of an audio signal  115 . Pre-processing of an audio signal can for example include filtering the audio signal  115  through a linear prediction filter, and/or perceptual weighting. In some implementations, the quantization, segmenting and/or any further pre-processing is performed in the encoder  110 , or such signal processing could have been performed in further equipment to which an input of the encoder  110  is connected. 
     In step  205 , a time-to-frequency transform is applied to the TD signal segment  T , so that a segment spectrum  S  is generated. The time-to-frequency transform could for example be a Discrete Fourier Transform (DFT), implemented e.g. as the Fast Fourier Transform: 
                       S   ⁡     (   k   )       =       ∑     n   =   0       N   -   1       ⁢       T   ⁡     (   n   )       ⁢     ⅇ     -       j   ⁢           ⁢   π   ⁢           ⁢   nk     N               ,           (   1   )               
where T(n) is a TD signal segment sample, nε[0, 1, . . . , N−1], and S(k) is the kth component of the complex DFT, kε[0, 1, . . . , N−1]
 
     Other possible transforms that could alternatively be used in step  205  include the discrete cosine transform, the Hadamard transform, the Karhunen-Loëve transform, the Singular Value Decomposition (SVD) transform, Quadrature Mirror Filter (QMF) filter banks, etc. Such transform algorithms are known in the art, and will not be further described here. 
     Step  205  typically includes determining the magnitude spectrum  X :
 
 X ( k )=| S ( k )|, k= 0,1,2,3  . . . M   (2),
 
where M=N/2+1 (assuming that N is even). If only the magnitude spectrum is required, it would hence be sufficient for k to run from k=0 to k=M, while if while if a full phase spectrum is desired, k would advantageously run from k=0 to k=N−1.
 
     In step  210 , the ASCB is searched for a vector which can provide a first approximation of the magnitude spectrum  X , and hence a first approximation of the segment spectrum  S . The ASCB can be seen as a matrix  C   A  having dimensions N ASCB ×M (or M×N ASCB ), where N ASCB  denotes the number of adaptive spectral code book vectors included in the ASCB, where a typical value of N ASCB  could lie within the range [16,128] (other values of N ASCB  could alternatively be used). Each row (or column) of the matrix  C   A  represents a synthesized magnitude spectrum of a previous segment, such that C A,i,k  (C A,k,i ) denotes frequency bin kε[0, 1, . . . , M−1] for segment m-i for i=1, 2, 3 . . . , N ASCB , where m denotes the current segment. For ease of description, it will in the following be assumed that the previous synthesized spectra are represented by the rows, rather than the columns, of the ASCB matrix  C   A . Furthermore, it will for illustrative purposes be assumed that the rows of  C   A  are normalized, such that: 
                   ∑     k   =   0       M   -   1       ⁢     C     A   ,   i   ,   k     2       =   1     ,     i   =   1     ,   2   ,     3   ⁢           ⁢   …     ⁢           ,     N   ASCB           
Normalization of the ASCB vectors stored in  C   A  will furthermore simplify the calculations.
 
     The search of the ASCB performed in step  210  could for example include determining the row vector of  C   A  which yields the largest absolute magnitude correlation with the segment spectrum: 
                       i   ASCB     =         arg   ⁢           ⁢   max     i     ⁢     (       ∑     k   =   0       M   -   1       ⁢       C     A   ,   i   ,   k       ⁢     X   ⁡     (   k   )           )         ,           (   3   )               
where i ASCB  is an index identifying the selected ASCB vector. Expression (3) can be seen as if the ASCB vector which matches the segment spectrum in a minimum mean squared error sense is selected. Other ways of selecting the ASCB vector may be employed, such as e.g. selecting the ASCB vector which minimizes the average error over a fixed number of consecutive segments.
 
     Once a row vector  C   A,i     ASCB    has been selected to provide an approximation of the magnitude spectrum  X , a gain parameter g ASCB  can be determined, for example by use of the following expression: 
                     g   ASCB     =       ∑     k   =   0       M   -   1       ⁢       C     A   ,     i   ASCB     ,   k       ⁢       X   ⁡     (   k   )       .                 (   4   )               
A first approximation of the segment spectrum can be given as g ASCB ·  C   A,i     ASCB   . Since  C   A,i     ASCB,k    and  X  are magnitude spectra, the gain g ASCB  will always be positive.
 
Step  215  is then entered, wherein the FSCB is searched for an FSCB vector providing an approximation of the residual spectrum, here referred to a residual spectrum approximation. The residual spectrum  R  can for example be defined as:
 
 R ( k )= X ( k )− g   ASCB   C   A,i     ASCB     ,k   k= 0,1,2, . . . ,( M− 1)  (5)
 
The FSCB can be seen as a matrix  C   F  having dimensions N FSCB ×M (or M×N FSCB ), where N FSCB  denotes the number of fixed spectral code book vectors included in the FSCB, where a typical value of N FSCB  could lie within the range [16,128] (other values of N FSCB  could alternatively be used). Each row (or column) of the matrix  C   F  represents a fixed differential spectrum, such that C F,i,k  (C F,k,i ) denotes frequency bin kε[0, 1, . . . , M−1] for entry number i=1, 2, 3 . . . , N FSCB . For ease of description, it will in the following be assumed that the previous synthesized spectra are represented by the rows, rather than the columns, of the FSCB matrix  C   F .
 
     The search of the FSCB performed in step  215  could for example include determining the row vector of  C   F  which yields the largest absolute magnitude correlation with the residual spectrum: 
                       i   FSCB     =         arg   ⁢           ⁢   max     i     ⁢     (       ∑     k   =   0       M   -   1       ⁢       C     F   ,   i   ,   k       ⁢     R   ⁡     (   k   )           )         ,           (   6   )               
where i FSCB  is an index identifying the selected FSCB vector to be used in providing the residual spectrum approximation.
 
     Once a row vector  C   F,i     FSCB    has been selected to provide an approximation of the residual spectrum, a gain parameter g FSCB  can be determined, for example by use of the following expression: 
     
       
         
           
             
               
                 
                   
                     g 
                     FSCB 
                   
                   = 
                   
                     
                       ∑ 
                       
                         k 
                         = 
                         0 
                       
                       
                         M 
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                     ⁢ 
                     
                       
                         C 
                         
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                           , 
                           
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     A residual spectrum approximation can be given as g FSCB ·  C   F,i     FSCB   . 
     A signal representation P of the signal segment is then generated in step  220 , the signal representation P being indicative of the indices i ASCB  and i FSCB , as well as of the gains g ASCB  and g FSCB . The representations of g ASCB  and g FSCB  included in the representation P are typically quantized, and could for example correspond to the values of g ASCB  &amp; g FSCB , or to the values of a global gain ratio 
                 g   ∝     =         g   FSCB       g   ASCB       ⁢     (       or   ⁢           ⁢     g   β       =       g   ASCB       g   FSCB         )         ,         
where the global gain represents the global energy of the signal segment. By representing the gains by (quantized values of) g ∝  and g global , the balance between energy matching and waveform matching can more easily be controlled, as described below in relation to expression (19). In the following, no difference will be made in the notation of actual gain values and the quantized gain values. Signal representation P forms part of the audio signal representation  120 .
 
     Step  225  is then entered, wherein the ASCB is updated with a vector  Y , or a vector proportional to  Y , where  Y  is the synthesized magnitude spectrum obtained from a linear combination of the selected ASCB vector  C   A,i     ASCB    and the selected FSCB vector  C   F,i     FSCB   :
 
 Y ′( k )= g   ASCB   C   A,i     ASCB     ,k   +g   FSCB   C   F,i     FSCB     ,k   (8a).
 
     In expression (8a), we assume that the synthesis is based on the gain parameter pair g ASCB  &amp; g FSCB . As mentioned above, the synthesis may be based on the gain parameter pair g global  and g α . The synthesized magnitude spectrum could then be expressed by:
 
 Y ′( k )= g   global ( C   A,i     ASCB     ,k   +g   α   C   F,i     FSCB     ,k )  (8b).
 
     Since the residual spectrum approximation is obtained as a differential spectrum, the FSCB gain can take a negative value. Furthermore, it may be that a simple linear combination of  C   A,i     ASCB    and  C   F,i     FSCB    yields negative values of the spectral magnitude for some frequency bins k. Hence, in order to obtain a physically correct representation of the synthesized segment spectrum, any negative frequency bin magnitude values could be replaced by zero, so that: 
     
       
         
           
             
               
                 
                   
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     Negative frequency bin magnitude values could alternatively be replaced by other positive values, such as |Y′|(k)|. 
     As will be seen below, it may in some implementations be beneficial to determine a pre-synthesis magnitude spectrum as:
 
 Y   pre ( k )= C   A,i     ASCB     ,k   +g   α   C   F,i     FSCB     ,k   (8d).
 
     Thus, the synthesized magnitude spectrum is determined in step  315  as  Y /g global , and the scaling with g global  is performed after the f-to-time transform. This is particularly useful if the synthesized TD signal segment is used for determining a suitable value of g global  (cf. expression (19) and (20)). 
     As mentioned above, order to simplify the numerical calculations illustrated by expressions (3) and (4) above, the rows of  C   A  can advantageously be normalized such that: 
     
       
         
           
             
               
                 
                   
                     
                       
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                   9 
                   ) 
                 
               
             
           
         
       
     
     In an implementation wherein the rows of  C   A  are normalized, the ASCB is hence updated with a normalized version of the magnitude spectrum  Y :
 
 C   A,U,k   :=Y   normalised ( k )
 
     where U denotes the row of ASCB to be updated, which typically is the row representing the oldest previous synthesized spectrum stored in the ASCB. An example of the updating procedure can be represented by first shifting the rows of the ASCB down one step such that:
 
 C   A,i,k   =C   A,i-1,k   ,i=N   ASCB , . . . ,4,3,2 k= 0,1,2, . . . ,( M− 1),  (10a)
 
and then, the normalized synthesized spectrum magnitude is inserted in the first row:
 
     
       
         
           
             
               
                 
                   
                     
                       C 
                       
                         A 
                         , 
                         1 
                         , 
                         k 
                       
                     
                     = 
                     
                       
                         Y 
                         ⁡ 
                         
                           ( 
                           k 
                           ) 
                         
                       
                       
                         
                           
                             ∑ 
                             
                               j 
                               = 
                               0 
                             
                             
                               M 
                               - 
                               1 
                             
                           
                           ⁢ 
                           
                             
                               Y 
                               2 
                             
                             ⁡ 
                             
                               ( 
                               j 
                               ) 
                             
                           
                         
                       
                     
                   
                   , 
                   
                     
 
                   
                   ⁢ 
                   
                     k 
                     = 
                     0 
                   
                   , 
                   1 
                   , 
                   2 
                   , 
                   
                     … 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     
                       ( 
                       
                         M 
                         - 
                         1 
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   
                     10 
                     ⁢ 
                     b 
                   
                   ) 
                 
               
             
           
         
       
     
     The ASCB could for example be implemented as a FIFO (First In First Out) buffer. From an implementation perspective, it is often advantageous to avoid the shifting operation of expressions (10a) &amp; (10b), and instead move the insertion point for the current frame, using the ASCB as a circular buffer. 
     Prior to having received any TD signal segments  T  to be encoded, the ASCB is preferably initialized in a suitable manner, for example by setting the elements of the matrix  C   A  to random numbers, or by using a pre-defined set of vectors. In one embodiment, here used as an example, the matrix  C   A  is initialized with a single constant value, corresponding to a set of flat spectra: 
     
       
         
           
             
               
                 
                   
                     C 
                     
                       A 
                       , 
                       i 
                       , 
                       k 
                     
                   
                   = 
                   
                     
                       1 
                       
                         M 
                       
                     
                     . 
                   
                 
               
               
                 
                   ( 
                   11 
                   ) 
                 
               
             
           
         
       
     
     The FSCB could for example be represented by a pre-trained vector codebook, which has the same structure as the ASCB, although it is not dynamically updated. There are several options for constructing an FSCB. An FSCB could for example be composed of a fixed set of differential spectrum candidates stored as vectors, or it could be generated by a number of pulses, as is commonly used in CELP coding for generation of time domain FCB vectors. Typically, a successful FSCB has the capability of introducing, into a synthesized segment spectrum (and hence into the ASCB), spectral components which have not been present in previous synthesized signals that represented in the ASCB. Pre-training of the FSCB could be performed using a large set of audio signals representing possible spectral magnitude distributions. 
     An encoder  110  could, if desired, as part of the encoding of a signal segment, furthermore generate a synthesized TD signal segment,  Z . This would correspond to performing step  320  of the decoding method flowchart illustrated in  FIG. 3 , and the encoder  110  could include corresponding TD signal segment synthesizing apparatus. The synthesis of the TD signal segment in the encoder  110 , as well as in the decoder  112 , could be beneficial if encoding parameters are determined in dependence of the synthesized TD signal segment, cf. for example expression (19) below. 
     An embodiment of a decoding method is shown in  FIG. 3 , which decoding method allows the decoding of a signal segment which has been encoded by means of the method illustrated in  FIG. 2 . At step  300 , a representation P of a signal segment is received in a decoder  112 . The representation P is indicative of an index i ASCB  &amp;an index i FSCB , a gain g ASCB  &amp; a gain g FSCB  (possibly represented by a global gain and a gain ratio). 
     At step  305 , a first ASCB vector  C   A,i     ASCB   , providing an approximation of the segment spectrum  S , is identified in an ASCB of the decoder  112  by means of the ASCB index i ASCB . The ASCB of the decoder  112  has the same structure as the ASCB of the encoder  110 , and has advantageously been initialized in the same manner. As will be seen in relation to step  325 , the ASCB of the decoder  112  is also updated in the same manner as the ASCB of the encoder  110 . At step  310 , an FSCB vector  C   F,i     FSCB    providing an approximation of the residual spectrum  R  is identified in an FSCB of the decoder  112  by means of the FSCB index i FSCB . The FSCB of the decoder  112  is advantageously identical to the FSCB of the encoder  110 , or, at least, comprises corresponding vectors  C   F,i     FSCB    which can be identified by FSCB indices i FSCB . 
     At step  315 , a synthesized magnitude spectrum  Y  is generated as a linear combination of the identified ASCB vector  C   A,i     ASCB    and the identified FSCB vector  C   F,i     FSCB   . Any negative frequency bin values are handled in the same manner as in step  225  of  FIG. 2  (cf. discussion in relation to expression (8)). 
     At step  320 , a frequency-to-time transform, i.e. the inverse of the time-to-frequency transform used in step  205  of  FIG. 2 , is applied to a synthesized spectrum  B  having the synthesized magnitude spectrum  Y  obtained in step  315 , resulting in a synthesized TD signal segment  Z . As will be further discussed below, a phase spectrum of the segment spectrum can also be taken into account when performing the inverse transform, for example as a random phase spectrum, or as a parameterized phase spectrum. Alternatively, a predetermined phase spectrum will be assumed for the synthesized spectrum  B . From the synthesized TD signal segment  Z , a synthesized audio signal  125  can be obtained. If any pre-processing had been performed in the encoder  110  prior to entering step  205 , the inverse of such pre-processing will be applied to the synthesized TD signal  Z  to obtain the synthesized audio signal  125 . 
     When the discrete Fourier transform (DFT) has been used by the encoder  110  in step  205 , the synthesized TD signal segment is obtained by applying, to the synthesized segment spectrum  B , the inverse DFT (IDFT): 
     
       
         
           
             
               
                 
                   
                     
                       Z 
                       ⁡ 
                       
                         ( 
                         n 
                         ) 
                       
                     
                     = 
                     
                       
                         1 
                         N 
                       
                       ⁢ 
                       
                         
                           ∑ 
                           
                             k 
                             = 
                             0 
                           
                           
                             N 
                             - 
                             1 
                           
                         
                         ⁢ 
                         
                           
                             B 
                             ⁡ 
                             
                               ( 
                               k 
                               ) 
                             
                           
                           ⁢ 
                           
                             ⅇ 
                             
                               j2π 
                               ⁢ 
                               
                                   
                               
                               ⁢ 
                               
                                 nk 
                                 / 
                                 N 
                               
                             
                           
                         
                       
                     
                   
                   , 
                   
                     
 
                   
                   ⁢ 
                   
                     n 
                     = 
                     0 
                   
                   , 
                   1 
                   , 
                   2 
                   , 
                   … 
                   ⁢ 
                   
                       
                   
                   , 
                   
                     
                       ( 
                       
                         N 
                         - 
                         1 
                       
                       ) 
                     
                     . 
                   
                 
               
               
                 
                   ( 
                   12 
                   ) 
                 
               
             
           
         
       
     
     When the discrete Fourier transform (DFT) is used for the encoding, step  320  could advantageously further include, prior to performing the IDFT, an operation whereby the symmetry of the DFT is reconstructed in order to obtain a real-valued signal in the time domain:
 
 B ( M+k )= B *( M−k ), k= 1,2,3, . . . ( M− 2)  (13)
 
where (*) denotes the complex conjugate operator.
 
     An encoder  110  which is configured to perform the method illustrated by  FIG. 2  is schematically shown in  FIG. 4 . The encoder  110  of  FIG. 4  comprises an input  400 , a t-to-f transformer  405 , an ASCB search unit  410 , an ASCB  415 , a residual spectrum generator  420 , an FSCB search unit  425 , an FSCB  430 , a magnitude spectrum synthesizer  435 , an index multiplexer  440  and an output  445 . Input  400  is arranged to receive a TD signal segment  T , and to forward the TD signal segment  T  the t-to-f transformer  405  to which it is connected. The t-to-f transformer  405  is arranged to apply a time-to-frequency transform to a received TD signal segment  T , as discussed above in relation to step  205  of  FIG. 2 , so that a segment spectrum  S  is obtained. The t-to-f transformer  405  of  FIG. 4  is further configured to derive the magnitude spectrum  X  of an obtained segment spectrum  S  by use of expression (2) above. The t-to-f transformer  405  of  FIG. 4  is connected to the ASCB search unit  410 , as well as to the residual spectrum generator  420 , and arranged to deliver a derived magnitude spectrum  X  to the ASCB search unit  410  as well as to the residual spectrum generator  420 . 
     The ASCB search unit  410  is further connected to the ASCB  415 , and configured to search for and select an ASCB vector  C   A,i     ASCB    which can provide a first approximation of the magnitude spectrum  X , for example using expression (3). The ASCB search unit  410  is further configured to deliver, to the index multiplexer  440 , a signal indicative of an ASCB index i ASCB  identifying the selected ASCB vector  C   A,i     ASCB   . The ASCB search unit  410  is further configured to determine a suitable ASCB gain, g ASCB , for example by use of expression (4) above, and to deliver, to the index multiplexer  440  as well as to the residual spectrum generator, a signal indicative of the determined ASCB gain g ASCB . The ASCB  415  is connected (for example responsively connected) to the ASCB search unit  410  and configured to deliver signals representing different ASCB vectors stored therein to the ASCB search unit  410  upon request from the ASCB search unit  410 . 
     The residual spectrum generator  420  is connected (for example responsively connected) to the ASCB search unit  410  and arranged to receive the selected ASCB vector  C   A,i     ASCB    and the ASCB gain from the ASCB search unit  410 . The residual spectrum generator  420  is configured to generate a residual spectrum  R  from a selected ASCB vector and gain received from the ASCB search unit  420 , and corresponding magnitude spectrum  X  received from the t-to-f transformer  420  (cf. expression (5). In the residual spectrum generator  420  of  FIG. 4 , an amplifier  421  and an adder  422  are provided for this purpose. The amplifier  421  is configured to receive the selected ASCB vector  C   A,i     ASCB    and the gain g ASCB , and to output a first approximation of the segment spectrum. The adder  422  is configured to receive the magnitude spectrum  X  as well as the first approximation of the segment spectrum; to subtract the first approximation from the magnitude spectrum  X ; and to output the resulting vector as the residual vector  R . 
     The FSCB search unit  425  is connected (for example responsively connected) to the output of residual spectrum generator  420  and configured to search for and select, in response to receipt of a residual spectrum  R , an FSCB vector  C   F,i     FSCB    which can provide a residual spectrum approximation, for example using expression (6). For this purpose, the FSCB search unit  425  is connected to the FSCB  430 , which is connected (for example responsively connected) to the FSCB search unit  425  and configured to deliver signals representing different FSCB vectors stored in FSCB  430  to the FSCB search unit  410  upon request from the FSCB search unit  410 . 
     The FSCB search unit  425  is further connected to the index multiplexer  440  and the spectrum magnitude synthesizer  435 , and configured to deliver, to the index multiplexer  440 , a signal indicative of an FSCB index i FSCB  identifying the selected FSCB vector  C   F,i     FSCB   . The FSCB search unit  425  is further configured to determine a suitable FSCB gain, g FSCB , for example by use of expression (7) above, and to deliver, to the index multiplexer  440  as well as to the spectrum magnitude synthesizer  435 , a signal indicative of the determined FSCB gain g FSCB . 
     The magnitude spectrum synthesizer  435  is connected (for example responsively connected) to the ASCB search unit  410  and the FSCB search unit  425 , and configured to generate a synthesized magnitude spectrum  Y . For this purpose, the magnitude spectrum synthesizer  435  of  FIG. 4  comprises two amplifiers  436  and  437 , as well as an adder  438 . Amplifier  436  is configured to receive the selected FSCB vector  C   F,i     FSCB    and the FSCB gain g FSCB  from the FSCB search unit  425 , while amplifier  437  is configured to receive the selected ASCB vector  C   A,i     ASCB    and the ASCB gain g ASCB  from the ASCB search unit  410 . Adder  438  is connected to the outputs of amplifier  436  and  437 , respectively, and configured to add the output signals, corresponding to the residual spectrum approximation and the first approximation of the segment spectrum, respectively, to form the synthesized magnitude spectrum  Y , which is delivered at an output of the magnitude spectrum synthesizer  435 . This output of the magnitude spectrum synthesizer  435  is connected to the ASCB  415 , so that the ASCB  415  may be updated with a synthesized magnitude spectrum  Y . The magnitude spectrum synthesizer  435  could further be configured to zero any frequency bins having a negative magnitude (cf. expression (8)), and/or to normalize the synthesized magnitude spectrum  Y  prior to delivering the synthesized spectrum  Y  to the ASCB  415 . Normalization of  Y  could alternatively be performed by the ASCB  415 , in a separate normalization unit connected between  435  and  415 , or be omitted. In an implementation wherein a synthesized TD signal segment is generated in the encoder  110 , the encoder  110  could furthermore advantageously include an f-to-t transformer connected to an output of the magnitude spectrum synthesizer  435  and configured to receive the (un-normalized) synthesized magnitude spectrum  Y . 
     As mentioned in the above, the index multiplexer  440  is connected to the ASCB search unit  410  and the FSCB search unit  425  so as to receive signals indicative of an ASCB index i ASCB  &amp; an FSCB index i FSCB , as well as an ASCB gain &amp; an FSCB index. The index multiplexer  440  is connected to the encoder output  445  and configured to generate a signal representation P, carrying a values indicative of an ASCB index i ASCB  &amp; an FSCB index i FSCB , as well as of a quantized values of the ASCB gain and the FSCB gain (or of a gain ratio and a global gain as discussed in relation to step  220  of  FIG. 2 ). 
       FIG. 5  is a schematic illustration of an example of a decoder  112  which is configured to decode a signal segment having been encoded by the encoder  110  of  FIG. 4 . The decoder  112  of  FIG. 5  comprises an input  500 , an index demultiplexer  505 , an ASCB identification unit  510 , an ASCB  515 , an FSCB identification unit  520 , an FSCB  525 , a magnitude spectrum synthesizer  530 , an f-to-t transformer  535  and an output  540 . The input  500  is configured to receive a signal representation P and to forward the signal representation P to the index demultiplexer  505 . The index demultiplexer  505  is configured to retrieve, from the signal representation P, values corresponding to an ASCB index i ASCB  &amp; an FSCB index i FSCB , and an ASCB gain g ASCB  &amp; an FSCB gain g FSCB  (or a global gain and a gain ratio). The index demultiplexer  505  is further connected to the ASCB identification unit  510 , the FSDC identification unit  520  and to the magnitude spectrum synthesizer  530 , and configured to deliver i ASCB  to the ASCB search unit  510 , to deliver i FSCB  to the FSCB search unit  520 , and to deliver g ASCB  as well as g FSCB  to the magnitude spectrum synthesizer  530 . 
     The ASCB identification unit  510  is connected (for example responsively connected) to the index demultiplexer  505  and arranged to identify, by means of a received value of the ASCB index i ASCB , an ASCB vector  C   A,i     ASCB    which was selected by the encoder  110  as the selected ASCB vector. The ASCB identification unit  510  is furthermore connected to the magnitude spectrum synthesizer  530 , and configured to deliver a signal indicative of the identified ASCB vector to the magnitude spectrum synthesizer  530 . Similarly, the FSCB identification unit  520  is responsibly connected to the index demultiplexer  505  and arranged to identify, by means of a received value of the FSCB index i ASCB , an FSCB vector  C   F,i     FSCB    which was selected by the encoder  110  as the selected FSCB vector. The FSCB identification unit  510  is furthermore connected to the magnitude spectrum synthesizer  530 , and configured to deliver a signal indicative of the identified FSCB vector to the magnitude spectrum synthesizer  530 . 
     The magnitude spectrum synthesizer  530  can, in one implementation, be identical to the magnitude spectrum synthesizer  435  of  FIG. 4 , and is shown to comprise an amplifier  531  configured to receive the identified ASCB vector  C   A,i     ASCB    &amp; the ASCB gain g ASCB , and an amplifier  532  configured to receive the identified FSCB vector  C   F,i     FSCB    &amp; the FSCB gain g FSCB . The adder  533  is configured to receive the output from the amplifier  531 , corresponding to the first approximation of the segment spectrum, as well as to receive the output from the amplifier  532 , corresponding to the residual spectrum approximation, and configured to add the two outputs in order to generate a synthesized magnitude spectrum  Y . The output of the magnitude spectrum synthesizer  530  is connected to the ASCB  515 , so that the ASCB  515  may be updated with a synthesized magnitude spectrum  Y . As the magnitude spectrum synthesizer  435 , the magnitude spectrum synthesizer  530  could further be configured to zero any frequency bins having a negative magnitude (cf. expression (8)), and/or to normalize the synthesized magnitude spectrum  Y  prior to delivering the synthesized spectrum  Y  to the ASCB  515 . Normalization of  Y  could alternatively be performed by the ASCB  515 , in a separate normalization unit connected between  530  and  515 , or be omitted, depending on whether or not normalization is performed in the encoder  110 . In any event, the magnitude spectrum synthesizer  435  is configured to deliver a signal indicative of the un-normalized synthesized magnitude spectrum  Y  to the f-to-t transformer  535 . 
     The f-to-t transformer  535  is connected (for example responsively connected) to the output of magnitude spectrum synthesizer  530 , and configured to receive a signal indicative of the synthesized magnitude spectrum  Y . The f-to-t transformer  535  is furthermore configured to apply, to a received synthesized magnitude spectrum  Y , the inverse of the time-to-frequency transform used in the encoder  110  (i.e. a frequency-to-time transform), in order to obtain a synthesized TD signal  Z . The f-to-t transformer  535  is connected to the decoder output  540 , and configured to deliver a synthesized TD signal to the output  540 . 
     In  FIGS. 4 and 5 , ASCB search unit  410  &amp; ASCB identification unit  510  are shown to be arranged to deliver a signal indicative of the selected/identified ASCB vector  C   A,i     ASCB   , while FSCB search unit  425  and FSCB identification unit  520  are similarly shown to be arranged to deliver a signal indicative of the selected/identified FSCB vector C F,i     FSCB   . In another implementation, the selected ASCB vector  C   A,i     ASCB    could be delivered directly from the ASCB  415 / 515 , upon request from the ASCB search unit  410 /ASCB identification unit  510 , and the selected FSCB vector  C   F,i     FSCB    could similarly be delivered directly from the FSCB  425 / 525 . 
     In  FIGS. 2-5 , the ASCB  415 / 515  is shown to be updated with the synthesized magnitude spectrum  Y . In one embodiment, this updating of the ASCB  415 / 515  is conditional on the properties of the synthesized magnitude spectrum  Y . A reason for providing a dynamic ASCB  415 / 515  is to adapt the possibilities of finding a suitable first approximation of a segment spectrum to a pattern in the audio signal  115  to be encoded. However, there may be some signal segments for which the segment spectrum  S  will not be particularly relevant to the encodability of any following signal segment. In order to allow for the ASCB  415 / 515  to include a larger number of useful ASCB vectors, a mechanism could be implemented which reduces the number of such irrelevant segment spectra introduced into the ASCB  415 / 515 . Examples of signal segments, for which the segment spectra could be considered irrelevant to the future encodability, are signal segments which are dominated by sounds that are not part of the content carrying audio signal that it is desired to encode, signal segments which are dominated by sounds that are not likely to be repeated; or signal segments which mainly carry silence or near-silence, etc. In the near-silence region, the synthesis would typically be sensitive to noise from numerical precision errors, and such spectra will be less useful for future predictions. 
     Hence, a check as to the relevance of a signal segment may be performed prior to updating the ASCB  415 / 15  with the corresponding synthesized magnitude spectrum  Y . An example of such check is illustrated in the flowchart of  FIG. 6 . The check of  FIG. 6  is applicable to both the encoder  110  and the decoder  112 , and if it has been implemented in one of them, it should be implemented in the other, in order to ensure that the ASCBs  415  and  515  include the same ASCB vectors. At step  600 , it is checked whether a signal segment m is relevant for the encodability of future signal segments. If so, step  225  (encoder) or step  325  (decoder) is entered, wherein the ASCB  415 / 515  is updated with the synthesized magnitude spectrum  Y   m . Step  200  (encoder) or step  300  (decoder) is then re-entered, wherein a signal representing the next signal segment m+1 is received. However, if it is found in step  600  that the signal segment m is irrelevant for the future encodability, then step  225 / 325  is omitted for segment m, and step  200 / 300  is re-entered without having performed step  225 / 325 . Step  600  could, if desired, be performed at an early stage in the encoding/decoding process, in which case several steps would typically be performed between step  600  and steps  225 / 325  or steps  200 / 300 . Although step  225 / 325  is shown in  FIG. 6  to be performed prior to the re-entering of the step  200 / 300 , there is no particular order in which these two steps should be performed. 
     In one implementation, the global energy g global  of the signal segment could be used as a relevance indicator. The check of step  600  could in this implementation be a check as to whether the global gain exceeds a global gain threshold: g global   m &gt;g global   threshold . If so, the ASCB  415 / 515  will be updated with  Y   m , otherwise not. In this implementation, the ASCB  415 / 515  will not be updated with spectra of signal segments which carry silence or near-silence, depending on how the threshold is set. 
     In another implementation, the encodability relevance check could involve a relevance classification of the content of signal segment. The relevance indicator could in this implementation be a parameter that takes one of two values: “relevant” or “not relevant”. For example, if the content of a signal segment is classified as “not relevant”, the updating of the ASCB  415 / 515  could be omitted for such signal segment. Relevance classification could for example be based on voice activity detection (VAD), whereby a signal segment is labeled as “voice active” or “voice inactive”. A voice inactive signal segment could be classified as “not relevant”, since its contents could be assumed to be less relevant to future encodability. VAD is known in the art and will not be discussed in detail. Relevance classification could for example be based on signal activity detection (SAD) as described in ITU-T G.718 section 6.2. A signal segment which is classified as active by means of SAD would be considered “relevant” for relevance classification purposes. 
     In an embodiment wherein the updating of the ASCB  415 / 515  is conditional on the relevance of a signal segment, the encoder  110  and decoder  112  will comprise a relevance checking unit, which could for example be connected to the output of the magnitude spectrum synthesizer  435 / 530 . An example of such relevance checking unit  700  is shown in  FIG. 7 . The relevance checking unit  700  is arranged to perform step  600  of  FIG. 6 . In one implementation, an analysis providing a value of a relevance indicator could be performed by the relevance checking unit  700  itself, or the relevance checking unit  700  could be provided with a value of a relevance indicator from another unit of the encoder  110 /decoder  112 , as indicated by the dashed line  705 . In  FIG. 7 , the relevance checking unit is shown to be connected to the magnitude spectrum synthesizer  435 / 530  and configured to receive a synthesized spectrum  Y   m . The relevance checking unit  700  is further arranged to perform the decision of step  600  of  FIG. 6 . For this decision, a value of a relevance indicator is typically required, as well as a value of a relevance threshold or a relevance fulfillment value. A relevance fulfillment value could for example be used instead of a relevance threshold if the relevance check involves a characterization of the content of the signal segment, the result of which can only take discrete values. The value of the relevance threshold/fulfillment value could advantageously be stored in the relevance checking unit  700 , for example in a data memory. Regarding the value of the relevance indicator, the relevance checking unit could, in one implementation, be configured to derive this value from  Y   m , for example if the relevance indicator is the global energy g energy . Alternatively, the relevance checking unit  700  could be configured to receive this value from another entity in the encoder  110 /decoder  112 , or be configured to receive a signal from which such value can be derived (e.g. a signal indicative of the TD signal segment  T ). The dashed arrow  705  in  FIG. 7  indicates that the relevance checking unit  700  may, in some embodiment, be connected to further entities from which signals can be received by means of which a value of the relevance parameter may be derived. The relevance checking unit  700  is further connected to the ASCB  415 / 515  and configured to, if the check of a signal segment indicates that the signal segment is relevant for the encodability of future signal segments, forward the synthesized magnitude spectrum  Y  to the ASCB  415 / 515 . 
     In some encoding situations, for example if the character of the audio signal  115  changes drastically so that the spectrum of a signal segment has few similarities with the spectra of previous signal segments, or when the ASCB  415 / 515  have just been initiated, there might not be an ASCB vector in the ASCB  415  which can provide a good approximation of the magnitude spectrum  X . In one embodiment, a fast convergence search mode of the codec is provided for such encoding situations. In the fast convergence search mode, a segment spectrum is synthesized by means of a linear combination of at least two FSCB vectors, instead of by means of a linear combination of one ASCB vector and one FSCB vector. In this mode, the bits allocated in the signal representation P for transmission of an ASCB index are instead used for the transmission of an additional FSCB index. Hence, the ASCB/FSCB bit allocation in the signal representation P is changed. 
     A criterion for entering into the fast convergence search mode could be that a quality estimate of the first approximation of the segment spectrum indicates that the quality of the first approximation would lie be below a quality threshold. An estimation of the quality of a first approximation could for example include identifying a first approximation of the segment spectrum by means of an ASCB search as described above, and then derive a quality measure (e.g. the ASCB gain, g ASCB ) and compare the derived quality measure to a quality measure threshold (e.g. a threshold ASCB gain, g ASCB   threshold ) A threshold ASCB gain could for example lie at 60 dB below nominal input level, or at a different level. The threshold ASCB gain is typically selected in dependence on the nominal input level. If the ASCB gain lies below the ASCB gain threshold, then the quality of the first approximation could be considered insufficient, and the fast convergence search mode could be entered. Alternatively, the quality estimation could be performed by means of an onset classification of the signal segment, prior to searching the ASCB  415 , where the onset classification is performed in a manner so as to detect rapid changes in the character of the audio signal  115 . If a change of the audio signal character between two segments lies above a change threshold, then the segment having the new character is classified as an onset segment. Hence, if an onset classification indicates that the segment is an onset segment, it can be assumed that the quality of the first approximation would be insufficient, had an ASCB search been performed, and no ASCB search would have to be carried out for the onset signal segment. Such onset classification could for example be based on detection of rapid changes of signal energy, on rapid changes of the spectral character of the audio signal  115 , or on rapid changes of any LP filter, if an LP filtering of the audio signal  115  is performed. Onset classification is known in the art, and will not be discussed in detail. 
       FIG. 8  is a flowchart schematically illustrating a method whereby the fast convergence search mode (FCM) can be entered. In step  800 , it is determined whether estimation as to the quality of the first approximation of the segment spectrum shows that the quality would be sufficient. If so, the encoder  110  will stay in normal operation, wherein an ASCB vector and an FSCB vector are used in the synthesis of the segment spectrum. However, if it is determined in step  800  that the quality of the first approximation will be insufficient, fast convergence search mode will be assumed, wherein a segment spectrum is synthesized by means of a linear combination of at least two FSCB vectors, instead of by means of a linear combination of one ASCB vector and one FSCB vector. In step  805 , a signal is sent to the FSCB search unit  425  to inform the FSCB search unit  425  that the fast convergence search mode should be applied to the current signal segment. Step  810  is also entered (and could, if desired, be performed before, or at the same time as, step  805 ), wherein a signal is sent to the index multiplexer  440 , informing the index multiplexer  440  that the fast convergence search mode should be signaled to the decoder  112 . The signal representation P could for example include a flag to be used for this purpose. 
     In an embodiment wherein the quality estimation is based on the evaluation of the ASCB gain, the ASCB search unit  415  of the encoder  110  could be equipped with a first approximation evaluation unit, which could for example be configured to operate according to the flowchart of  FIG. 8 , where step  800  could involve a comparison of the ASCB gain to the threshold ASCB gain. In an embodiment wherein the quality estimation is based on a detection of rapid changes in the audio signal  115 , an onset classifier could be provided, either in the encoder  110 , or in equipment external to the encoder  110 . 
     In the fast convergence search mode, the FSCB code book is in step  215  searched for at least two FSCB vectors instead of one. In one implementation, wherein the FSCB code book is searched for two FSCB vectors in the FCM, an index pair (i FCB,1 ,i FCB,2 ) is desired which minimizes the error given by the following expression: 
     
       
         
           
             
               
                 
                   
                     ( 
                     
                       
                         i 
                         
                           FSCB 
                           , 
                           1 
                         
                       
                       , 
                       
                         i 
                         
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                           , 
                           2 
                         
                       
                     
                     ) 
                   
                   = 
                   
                     
                       
                         arg 
                         ⁢ 
                         
                             
                         
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                         ( 
                         
                           
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     The two FSCB gains can, just like the gains in the normal mode, be described by means of a global energy g energy  and a gain ratio, 
     
       
         
           
             
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     In an embodiment wherein the fast convergence search mode is provided as an alternative to normal encoding, the FSCB search unit  425  of the decoder could advantageously be connected to the magnitude spectrum synthesizer  435  in a manner so that the FSCB search unit can, when in fast convergence search mode, provide input signals to the amplifier  437 , as well as to the amplifier  436 . The spectral synthesis in the fast convergence search mode can be described by:
 
 Y ′( k )= g   FSCB,1   C   F,i     FSCB,1     ,k   +g   FSCB,2   C   F,i     FSCB,2     ,k   (15a),
 
or
 
 Y ′( k )= g   global ( C   F,i     FSCB,1     ,k   +g   α   C   F,i     FSCB,2     ,k ).  (15b).
 
     In the decoder, the index de-multiplexer  505  should advantageously be configured to determine whether an FCM indication is present in the signal representation P, and if so, to send the two vector indices of the signal representation P to the FSCB identification unit  520  (possibly together with an indication that the fast convergence search mode should be applied). The FSCB identification unit  520  is, in this embodiment, configured to identify two FSCB vectors in the FSCB  525  upon the receipt of two FSCB indices in respect of the same signal segment. The FSCB identification unit  520  is further advantageously connected to the magnitude spectrum synthesizer  530  in a manner so that the FSCB identification unit  530  can, when in fast convergence search mode, provide input signals to the amplifier  431 , as well as to the amplifier  532 . 
     The fast convergence search mode could be applied on a segment-by-segment basis, or the encoder  110  and decoder  112  could be configured to apply the FCM to a set of n consecutive signal segments once the FCM has been initiated. The updating of the ASCB  415 / 515  with the synthesized magnitude spectrum can in the fast convergence search mode advantageously be performed in the same manner as in the normal mode. 
     As discussed above, a synthesized segment spectrum  B  is obtained from a synthesized magnitude spectrum  Y , and the above description concerns the encoding of the magnitude spectrum  X  of a segment spectrum. However, audio signals are also sensitive to the phase of the spectrum. Hence, the phase spectrum of a signal segment could also be determined and encoded in the encoding method of  FIG. 2 . The representation of the segment spectrum  S  would then be divided into the magnitude spectrum  X  and a phase spectrum  φ :
 
 X ( k )=| S ( k )|, k= 0,1,2,3 . . . ( M− 1)  (16a)
 
Φ( k )=∠ S ( k ), k= 0,1,2,3 . . . ( M− 1)  (16b)
 
     The t-to-f transformer  405  could be configured to determine the phase spectrum. A phase encoder could, in one embodiment, be included in the encoder  110 , where the phase encoder is configured to encode the phase spectrum and to deliver a signal indicative of the encoded phase spectrum to the index multiplexer  440 , to be included in the signal representation P to be transmitted to the decoder  112 . The parameterization of the phase spectrum  φ  could for example be performed in accordance with the method described in section 3.2 of “ High Quality Coding of Wideband Audio Signals using Transform Coded Excitation  ( TCX )”, R. Lefebvre et al.,  ICASSP  1994, pp. I/193-I/196 vol. 1, or by any other suitable method. A synthesized segment spectrum  B  will take the form:
 
 B ( k )= Y ( k )· e   j2π·φ(k)   ,k= 1,2,3 . . . ,( M− 2)  (17).
 
     The DC component of B (k=0) and the Nyquist frequency component (k=M−1) are real values. 
     However, for signal segments carrying noise-like audio information, such as fricatives, the phase spectrum is generally not as important as for signal segments carrying harmonic content, such as voiced sounds or music. 
     For a phase insensitive signal segment, which could for example be a signal segment carrying noise or noise-like sounds (e.g. unvoiced sounds), the full phase spectrum  φ  does not have to be determined and parameterized. Hence, less information will have to be transmitted to the decoder  112 , and bandwidth can be saved. However, to base the synthesized segment spectrum on the synthesized magnitude spectrum only, and thereby use the same phase spectrum for all segment spectra, will typically introduce undesired artefacts. By assigning a random, or pseudo-random, phase spectrum to the synthesized segment spectrum  B , such undesired artefacts can much be avoided. The random phase spectrum is here denoted  V . The final complex synthesized phase spectrum would then be: 
                     B   ⁡     (   k   )       =     {             Y   ⁡     (   k   )       ,           k   =   0                   Y   ⁡     (   k   )       ·     ⅇ     j2π   ·   V         ,             k   =   1     ,   2   ,     3   ⁢           ⁢   …     ⁢           ,     (     M   -   2     )                   Y   ⁡     (   k   )       ,             k   =     M   -   1       ,                     (   18   )               
where V(k) represents a pseudo-random variable which can advantageously have a uniform distribution in the range [0,1]. Therefore, the phase information provided to the f-to-t transformer  535  of the decoder  112  (or to a corresponding f-to-t-transformer of the encoder  110 ) in relation to phase insensitive segments could be based on information generated by a random generator in the decoder  112 . The decoder  112  could, for this purpose, for example include a deterministic pseudo-random generator providing values having a uniform distribution in the range [0,1]. Such deterministic pseudo-random generators are well known in the art and will not be further described. Similarly, in applications wherein the encoder  110  is also configured to generate the full synthesized complex segment spectrum  B , in addition to the synthesized magnitude spectrum  Y , the encoder  110  could include such pseudo-random generator. In order for the encoder  110  and the decoder  112  to be synchronized, the same seed could advantageously be provided, in relation to the same signal segment, to the pseudo-random generators of the encoder  110  and the decoder  112 . The seed could e.g. be pre-determined and stored in the encoder  110  and decoder  112 , or the seed could be obtained from the contents of a specified part of the signal representation P upon the start of a communications session. If desired, the synchronization of random phase generation between the encoder  110  and decoder  112  could be repeated at regular intervals, e.g. 10 th  or 100 th  frame, in order to ensure that the encoder and decoder syntheses remain in synchronization.
 
     In one implementation of an encoding mode wherein a random phase spectrum  V  is used in the generation of a synthesized segment spectrum  B  for phase insensitive segments, the sign of the real valued component of the segment spectrum  S  is determined and signaled to the decoder  112 , in order for the decoder  112  to be able to use the sign of the DC component in the generation of  B . Adjusting the sign of the DC component of the synthesized segment spectrum  B  improves the stability of the energy evolution between adjacent segments. This is particularly beneficial in implementations where the segment length is short (for example in the order of 5 ms). When the segment length is short, the DC component will be affected by the local waveform fluctuations. By encoding the sign of the DC component as part of the signal representation P, sharp transitions at the segment boundaries, which otherwise may be present when a random phase spectrum is used, can generally be avoided. To provide information to the decoder  112  on the sign of the DC component of the phase spectrum, but to let the remaining parts of the phase spectrum used in the generation of the synthesized TD signal segment Z be randomly generated, can be seen as if one region (namely the DC component) of the phase spectrum is treated as phase sensitive, whereas another region (namely all other frequency components) are treated as phase insensitive. 
     At the decoder side, information on the phase spectrum  φ  will be taken into account in step  320 , wherein the f-to-t transform is applied to the synthesized spectrum. The f-to-t transformer  535  of  FIG. 5  could advantageously be connected to the index de-multiplexer  505  (as well as to the output of the magnitude spectrum synthesizer  530 ) and configured to receive a signal indicative of information on the phase spectrum  φ  of the segment spectrum, where such information is present in the signal representation P. Alternatively, the generation of a synthesized spectrum from a synthesized magnitude spectrum and received phase information could be performed in a separate spectrum synthesis unit, the output which is connected to the f-to-t transformer  530 . As discussed above, phase information included in P could for example be a full parameterization of a phase spectrum, or a sign of the DC component of the phase spectrum. Furthermore, when a random phase spectrum is used at least for some signal segments, the f-to-t transformer  535  (or a separate spectrum synthesis unit) could be connected to a random phase generator. 
       FIG. 9  schematically illustrates an example of an encoder  110  configured to provide an encoded signal P to a decoder  112  wherein a random phase spectrum  V , as well as information on the sign of the DC component, is used in generation of the synthesized TD signal segment  Z . Only mechanisms relevant to the phase aspect of the encoding have been included in  FIG. 9 , and the decoder  110  typically further includes other mechanisms shown in  FIG. 5 . In the embodiment of  FIG. 9 , the encoder  110  comprises a DC encoder  900 , which is connected (for example responsively connected) to the t-to-f transformer  405  and configured to receive a segment spectrum  S  from the transformer  405 . The DC encoder  900  is further configured to determine the sign of the DC component of the segment spectrum, and to send a signal DC± indicative of this sign to the index multiplexer  440 , which is configured to include an indication of the DC sign in the signal representation P, for example as a flag indicator. 
     In an embodiment wherein a full parameterized phase spectrum is included in the signal representation P, the DC encoder  900  could be replaced or supplemented with a phase encoder configured to parameterize the full phase spectrum. In another embodiment, values representing the phase of some, but not all, frequency bins are parameterized, for example the p first frequency bins, p&lt;N. 
       FIG. 10  schematically illustrates an example of a decoder  112  capable of decoding a signal representation P generated by the encoder  110  of  FIG. 9 . The decoder  112  of  FIG. 10  comprises, in addition to the mechanisms shown in  FIG. 5 , a random phase generator  1000  connected to the f-to-t transformer  535  and configured to generate, and deliver to transformer  535 , a pseudo-random phase spectrum  V  as discussed in relation to expression (18). In the embodiment of  FIG. 10 , the f-to-t transformer  535  is further configured to receive, from the index de-multiplexer  505 , a signal indicative of the sign of the DC component of a segment spectrum, in addition to being configured to receive a synthesized magnitude spectrum  Y . The transformer  535  is configured to generate a synthesized TD signal segment  Z  in accordance with the received information (cf. expression (18)). 
     In an implementation of the encoder  110  wherein the synthesized TD signal segment  Z  is generated in the encoder  110 , the encoder  110  would include a random phase generator  1000  and a f-to-t transformer  535  as shown in  FIG. 10 . 
     In an embodiment wherein a full parameterized phase spectrum is included in the signal representation P, the f-to-t transformer  535  of  FIG. 10  could be configured to receive a signal of this parameterized phase spectrum from the index de-multiplexer  505 . In an implementation wherein such information is provided for all signal segments, the random phase generator could be omitted. 
     In one embodiment, a signal segment is classified as either “phase sensitive” or “phase insensitive”, and the encoding mode used in the encoding of the signal segment will depend on the result of the phase sensitivity classification. In this embodiment, the encoder  110  has a phase sensitive encoding mode and a phase insensitive encoding mode, while the decoder  112  has a phase sensitive decoding mode as well as a phase insensitive decoding mode. Such phase sensitivity classification could be performed in the time domain, prior to the f-to-t transform being applied to the TD signal segment  T  (e.g. at a pre-processing stage prior to the signal having reached the encoder  110 , or in the encoder  110 ). Phase sensitivity classification could for example be based on a Zero Crossing Rate (ZCR) analysis, where a high rate of zero crossings of the signal magnitude indicates phase insensitivity—if the ZCR of a signal segment lies above a ZCR threshold, the signal segment would be classified as phase insensitive. ZCR analysis as such is known in the art and will not be discussed in detail. Phase sensitivity classification could alternatively, or in addition to an ZCR analysis, be based on spectral tilt—a positive spectral tilt typically indicates a fricative sound, and hence phase insensitivity. Spectral tilt analysis as such is also known in the art. Phase sensitivity classification could for example be performed along the lines of the signal type classifier described in ITU-T G.718, section 7.7.2. 
     A schematic flowchart illustrating an example of such classification is shown in  FIG. 11 . The classification could be performed in a segment classifier, which could form part of the encoder  110 , or be included in a part of the user equipment  105  which is external to the encoder  110 . In step  1100 , a signal indicative of a signal segment is received by a segment classifier, such as the TD signal segment  T , a signal representing the signal segment prior to any pre-processing, or a signal representing the segment spectrum,  S  or  X . At step  905 , it is determined whether the signal segment is phase insensitive. If so, the phase insensitive mode is entered in step  1110 . If not, the phase sensitive mode is entered step  1115 . In this embodiment, the phase insensitive mode is a transform-based adaptive encoding mode wherein a random phase spectrum  V  is used in the generation of the synthesized spectrum, possibly in combination with information on the sign of the DC component of the segment spectrum  S , or information on the phase value of a few of the frequency bins, as described above. The phase sensitive encoding mode can for example be a time domain based encoding method, wherein the TD signal segment  T  does not undergo any time-to-frequency transform, and where the encoding does not involve the encoding of the segment spectrum. For example, the phase sensitive encoding mode could involve encoding by means of a CELP encoding method. Alternatively, the phase sensitive encoding mode can be a transform based adaptive encoding mode wherein a parameterization of the phase spectrum is signaled to the decoder  112  instead of using a random phase spectrum  V . 
     Information indicative of which encoding mode has been applied to a particular segment could advantageously be included in the signal representation P, for example by means of a flag, so that the decoder  110  will be aware of which decoding mode to apply. 
     The encoding of phase information relating to a phase insensitive signal segment can, as seen above, be made by use of fewer bits than the encoding of a the phase information of a phase sensitive signal. In an implementation wherein the phase sensitive mode is also a transform based encoding mode, the encoding of a phase insensitive signal segment could be performed such that the bits saved from the phase quantization are used for improving the overall quality, e.g. by using enhanced temporal shaping in noise-like segments. 
     The encoding mode wherein a random phase spectrum  V  is used in the generation of a synthesized segment spectrum  B  is typically beneficial for both background noises and noise-like active speech segments such as fricatives. One characteristic difference between these sound classes is the spectral tilt, which often has a pronounced upward slope for active speech segments, while the spectral tilt of background noise typically exhibits little or no slope. The spectral modeling can be simplified by compensating for the spectral tilt in a known manner in case of active speech segments. For this purpose, a voice activity detector (VAD) could be included in the encoding user equipment  105   a , arranged to analyze signal segments in a known manner to detect active speech. The encoder  110  could include a spectral tilt mechanism, configured to apply a suitable tilt to a TD signal segment  T  in case active speech has been detected. A VAD flag could be included in the signal representation P, and the detector  112  could be provided with an inverse spectral tilt mechanism which would apply the inverse spectral tilt in a known manner to the synthesized TD signal segment  Z  in case the VAD flag indicates active speech. For audio signals that show strong variation in the spectral tilt, this tilt compensation simplifies the spectral modeling following ASCB and FSCB searches. 
     In an implementation wherein two different encoding modes are available, and wherein different signal segments can be encoded by either one of the encoding modes, waveform and energy matching between the two encoding modes might be desirable to provide smooth transitions between the encoding modes. A switch of signal modeling and of error minimization criteria may give abrupt and perceptually annoying changes in energy, which can be reduced by such waveform and energy matching. Waveform and energy matching can for instance be beneficial when one encoding mode is a waveform matching time domain encoding mode and the other is a spectrum matching transform based encoding mode, or when two different transform based encoding modes are used. For this purpose, the following expression for the global gain g global  could provide a balance between the energy and waveform matching: 
                       g   global     =       β   ⁢           ∑     n   =   0       N   -   1       ⁢       T   2     ⁡     (   n   )                 ∑     n   =   o       N   -   1       ⁢       Z   2     ⁡     (   n   )               +       (     1   -   β     )     ⁢           ∑     n   =   0       N   -   1       ⁢       r   ⁡     (   n   )       ⁢     Z   ⁡     (   n   )                   ∑     n   =   0       N   -   1       ⁢       Z   2     ⁡     (   n   )                   ,           (   19   )               
where the first term represents the contribution to the global gain from the matching of energies between the two encoding modes, the second term represents the contribution from the waveform matching, and β is a parameter βε[0,1] by which the balance between waveform and energy matching can be tuned. In one implementation, β is adaptive to the properties of the signal segment. The possibility of tuning the balance between waveform and energy matching is particularly useful when the encoding of an audio signal can be performed in two different encoding modes, such that an energy step may occur in transitions between the encoding modes. When one available encoding mode is a phase insensitive encoding mode as discussed above wherein at least part of the phase information is random, and the other encoding mode is a CELP based encoding method, a suitable value of β for encoding of a phase insensitive segment may for example lie in the range of [0.5,0.9], e.g. 0.7, which gives a reasonable energy matching while keeping smooth transitions between phase sensitive (e.g. voiced) and phase insensitive (e.g. unvoiced) segments. Other values of β may alternatively be used. In a case where most of the synthesized phase information is random, the second term of the expression for g global  will typically be close to zero and could be neglected. So for the case of all-random phase, the expression in (19) can be simplified to a constant attenuation of the signal energy using the constant factor β. Such energy attenuation reflects that the spectrum matching typically yields a better match and hence higher energy than the CELP mode on noise-like segments, and the attenuation serves to even out this energy difference for smoother switching.
 
     The global gain parameter g global  is typically quantized to be used by the decoder  112  to scale the decoded signal (for example when determining the synthesized magnitude spectrum according to expressions (8b) or (15b), or, by scaling the synthesized TD signal segment  Z  if, in step  315 , the synthesized segment spectrum is determined as  Y   pre .) 
     In an implementation wherein only one encoding mode is available for the encoding of a signal segment, a value of the global gain could for example be determined according to the following expression: 
     
       
         
           
             
               
                 
                   
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     As mentioned above, the TD signal segment  T  could have been pre-processed prior to entering the encoder  110  (or in another part of the encoder  110 , not shown in  FIG. 4 ). Such pre-processing could for example include perceptual weighting of the TD signal segment in a known manner. Perceptual weighting could, as an alternative or in addition to perceptual weighting prior to the t-to-f transform, be applied after the t-to-f transform of step  205 . A corresponding inverse perceptual weighting step would then be performed in the decoder  112  prior to applying the f-to-t transform in step  320 . A flowchart illustrating a method to be performed in an encoder  110  providing perceptual weighting is shown in  FIG. 12 . The encoding method of  FIG. 12  comprises a perceptual weighting step  1200  which is performed prior to the t-to-f transform step  205 . Here, the TD signal segment  T  is transformed to a perceptual domain where the signal properties are emphasized or de-emphasized to correspond to human auditory perception. This step can be made adaptive to the input signal, in which case the parameters of the transformation may need to be encoded to be used by the decoder  112  in a reversed transformation. The perceptual transformation may include one or several steps, e.g. changing the spectral shape of the signal by means of a perceptual filter or changing the frequency resolution by applying frequency warping. Perceptual weighting in known in the art, and will not be discussed in detail. A further, pre-coding weighting step is provided in step  1205 , which is entered after the t-to-f transform step  205 , prior to the ASCB search in step  220 . Both step  1200  and step  1205  are optional—one of them could be included, but not the other, or both, or none of them. Perceptual weighting could also be performed in an optional LP filtering step (not shown). Hence, the perceptual weighting could be applied in combination with an LP-filter, or on its own. 
     A flowchart illustrating a corresponding method to be performed in a decoder  110  providing perceptual weighting is shown in  FIG. 13 . The decoding method of  FIG. 13  comprises an inverse pre-coding weighting step  1300  which is performed prior to the f-to-t transform step  320 . Here, the synthesized signal spectrum magnitude  Y  is transformed to a perceptual domain where the signal properties are emphasized or de-emphasized to correspond to human auditory perception. The method of  FIG. 13  further comprises an inverse perceptual weighting step  1305 , performed after the f-to-t transform step  320 . If the encoding method includes step  1200 , then the decoding method includes step  1305 , and if the encoding method includes step  1205 , then the decoding method includes step  1300 . The application of perceptual weighting will not affect the general method, but will affect which ASCB vectors and FSCB vectors will be selected in steps  210  and  215  of  FIG. 2 . Preferably, the training of the FSCB  430 / 525  should take any weighting into account, so that the FSCB  430 / 525  includes FSCB vectors suitable for an encoding method employing perceptual weighting. 
     In  FIGS. 14-16 , two different examples of implementations of the above described technology are shown. 
     In  FIG. 14 , an example of an implementation of an encoder  110  wherein conditional updating, spectral tilting in dependence on VAD, DC sign encoding, random phase complex spectrum generation and mixed energy and waveform matching is performed on a LP filtered TD signal segment  T  is shown. The signals E(k) and E 2 (k) indicate signals to be minimized in the ASCB search and FSCB search, respectively (cf. expressions (3) and (6), respectively). Reference numerals  1 - 6  indicating the origin of different parameters to be included in the signal representation P, where the reference numerals indicate the following parameters:  1 : i ASCB ;  2 : g ASCB ;  3 : i FSCB ;  4 : g FSCB ;  5 : DC±;  6 : g global . 
     In  FIG. 15 , a corresponding decoder  112  is schematically illustrated. 
       FIG. 16  schematically illustrates an implementation of an encoder  110  wherein phase encoding, pre-coding weighting and energy matching is performed. A perceptual weight W(k) is derived from the TD signal segment T(n) and the magnitude spectrum X(k), and is taken into account in the ASCB search, as well as in the FSCB search, so that signals E w (k) and E w2 (k) are signals to be minimized in the ASCB search and FSCB search, respectively. The energy matching could for example be performed in accordance with expression (20). The encoder  110  of  FIG. 16  does not provide any local synthesis. In  FIG. 16 , reference numerals  1 - 6  indicate the following parameters:  1 : i ASCB ,  2 : g ASCB ;  3 : i FSCB ,  4 : g FSCB ;  5 : φ(k);  6 : g global . Here, explicit values of g ASCB  and g FSCB  are included in P together with a value of g global , instead of a value of g global  and the gain ratio g α , as in the implementation shown in  FIG. 14 . 
     The encoder of  FIG. 16  is configured to include values of g ASCB  &amp; g FSCB , as well as a value of g global  in the signal representation P, while the encoder of  FIG. 14  is configured to include a value of the gain ratio and a value of the global gain in P. 
       FIG. 17  schematically illustrates a decoder  112  arranged to decode a signal representation P received from the encoder  110 . 
     The encoder  110  and the decoder  112  could be implemented by use of a suitable combination of hardware and software. In  FIG. 18 , an alternative way of schematically illustrating an encoder  110  is shown (cf.  FIGS. 4 ,  14  and  16 ).  FIG. 18  shows the encoder  110  comprising a processor  1800  connected to a memory  1805 , as well as to input  400  and output  445 . The memory  1805  comprises computer readable means that stores computer program(s)  1810 , which when executed by the processing means  1800  causes the encoder  110  to perform the method illustrated in  FIG. 2  (or an embodiment thereof). In other words, the encoder  110  and its mechanisms  405 ,  410 ,  420 ,  425 ,  435  and  440  may in this embodiment be implemented with the help of corresponding program modules of the computer program  1810 . Processor  1800  is further connected to a data buffer  1815 , whereby the ASCB  415  is implemented. FSCB  430  is implemented as part of memory  1805 , such part for example being a separate memory. An FSCB  525  could for example be stored in a RWM (Read-Write) memory or ROM (Read-Only) memory. 
     The illustration of  FIG. 18  could alternatively represent an alternative way of illustrating a decoder  112  (cf.  FIGS. 5 ,  15  and  17 ), wherein the decoder  112  comprises a processor  1800 , a memory  1805  that stores computer program(s)  1810 , which, when executed by the processing means  1800  causes the decoder  112  to perform the method illustrated in  FIG. 3  (or an embodiment thereof). In this representation of the decoder, ASCB  515  is implemented by means of data buffer  1815 , and FSCB  525  is implemented as part of memory  1805 . Hence, the decoder  110  and its mechanisms  505 ,  510 ,  520 ,  530  and  535  may in this embodiment be implemented with the help of corresponding program modules of the computer program  1810 . 
     The processor  1800  could, in an implementation, be one or more physical processors—for example, in the encoder case, one physical processor could be arranged to execute code relating to the t-to-f transform, and another processor could be employed in the ASCB search, etc. The processor could be a single CPU (Central processing unit), or it could comprise two or more processing units. For example, the processor may include general purpose microprocessors, instruction set processors and/or related chips sets and/or special purpose microprocessors such as ASICs (Application Specific Integrated Circuit). The processor may also comprise board memory for caching purposes. 
     Memory  1805  comprises a computer readable medium on which the computer program modules, as well as the FSCB  525 , are stored. The memory  1805  could be any type of non-volatile computer readable memories, such as a hard drive, a flash memory, a CD, a DVD, an EEPROM etc, or a combination of different computer readable memories. The computer program modules described above could in alternative embodiments be distributed on different computer program products in the form of memories within an encoder  110 /decoder  112 . The buffer  1815  is configured to hold a dynamically updated ASCB  415 / 515  and could be any type of read/write memory with fast access. In one implementation, the buffer  1815  forms part of memory  1805 . 
     For purposes of illustration only, the above description has been made in terms of the frequency domain representation of a time domain signal segment being a segment spectrum obtained by applying a time-to-frequency transform to the signal segment. However, other ways of obtaining a frequency domain representation of a signal segment may be employed, such as a Linear Prediction (LP) analysis, a Modified Discrete Cosine Transform analysis, or any other frequency analysis, where the term frequency analysis here refers to an analysis which, when performed on a time domain signal segment, yields a frequency domain representation of the signal segment. A typical LP analysis includes calculating of the short-term auto-correlation function from the time domain signal segment and obtaining LP coefficients of an LP filter using the well-known Levinson-Durbin recursion. Examples of an LP analysis and the corresponding time domain synthesis can be found in references describing CELP codecs, e.g. ITU-T G.718 section 6.4. An example of a suitable MDCT analysis and the corresponding time domain synthesis can for example be found in ITU-T G.718 sections 6.11.2 and 7.10.6. 
     In an implementation wherein another frequency analysis than a time-to-frequency transform is employed, step  205  of the encoding method would be replaced by a step wherein another frequency analysis is performed, yielding another frequency domain representation. Similarly, step  305  would be replaced by a corresponding time domain synthesis based on the frequency domain representation. The remaining steps of the encoding method and decoding method could be performed in accordance with the description given in relation to using a time-to-frequency transform. An ASCB  415  is searched for an ASCB vector providing a first approximation of the frequency domain representation; a residual frequency representation is generated as the difference between the frequency domain representation and the selected ASCB vector, and an FSCB  425  is searched for an FSCB vector which provides an approximation of the residual frequency representation. However, the contents of the FSCBs  425 / 525 , and hence the contents of the ASCB  415 / 515 , could advantageously be adapted to the employed frequency analysis. The result of an LP analysis will be an LP filter. In an implementation wherein the frequency domain representation of a signal segment is obtained by use of an LP analysis, the ASCBs  415 / 515  would comprise ASCB vectors which could provide an approximation of the LP filter obtained from performing the LP analysis on a signal segment, and the FSCBs  425 / 525  would comprise FSCB vectors representing differential LP filter candidates, in a manner corresponding to that described above in relation to a frequency domain representation obtained by use of a time-to-frequency transform. Similarly, in an implementation wherein the frequency domain representation of a signal segment is obtained by performing an MDCT analysis on the signal segment, the ASCBs  415 / 515  would comprise ASCB vectors which could provide an approximation of an MDCT spectrum obtained from performing the MDCT analysis on a signal segment, and the FSCBs  425 / 525  could comprise FSCB vectors representing differential MDCT spectrum candidates. 
     When an LP analysis is used as the frequency analysis, the LP filter coefficients obtained from the LP analysis could, if desired, be converted from prediction coefficients to a domain which is more robust for approximations, such as for example an immitance spectral pairs (ISP) domain, (see for example ITU-T G.718 section 6.4.4). Other examples of suitable domains are a Line Spectral Frequency domain (LSF), an Immitance Spectral Frequency (ISF) domain or the Line Spectral Pairs (LSP) domain. Since small approximations on the LP coefficients themselves may lead to a large degradation in the performance of the LP filter, it is often advantageous to perform such conversion of the coefficients into a more robust domain, and the converted representation is used for quantization and interpolation of the LP filter. 
     The LP filter would in this implementation not provide a phase representation, but the LP filter could be complemented with a time domain excitation signal, representing an approximation of the LP residual. For phase insensitive segments, the time domain excitation signal could be generated with a random generator. For phase sensitive segments, the time domain excitation signal could be encoded with any type of time or frequency domain waveform encoding, e.g. the pulse excitation used in CELP, PCM, ADPCM, MDCT-coding etc. The generation of a synthesized TD signal segment (corresponding to step  320  of  FIGS. 3 and 13 ) from the frequency domain representation would in this case be performed by filtering the time domain excitation signal through the frequency domain representation LP filter. 
     The above described invention can be for example be applied to the encoding of audio signals in a communications network in both fixed and mobile communications services used for both point-to-point calls or teleconferencing scenarios. In such systems, a user equipment could be equipped with an encoder  110  and/or a decoder  112  as described above. The invention is however also applicable to other audio encoding scenarios, such as audio streaming applications and audio storage. 
     The advantages of the described technology in terms of improved encoding of noise-like sounds such as fricatives are particularly significant at low bitrates, since it is at the low bit rates that the known encoding methods are particularly weak. However, the technology described herein is applicable to audio encoding at any bit rate. 
     Although various aspects of the invention are set out in the accompanying independent claims, other aspects of the invention include the combination of any features presented in the above description and/or in the accompanying claims, and not solely the combinations explicitly set out in the accompanying claims. 
     One skilled in the art will appreciate that the technology presented herein is not limited to the embodiments disclosed in the accompanying drawings and the foregoing detailed description, which are presented for purposes of illustration only, but it can be implemented in a number of different ways, and it is defined by the following claims.