Patent Publication Number: US-7595753-B2

Title: Broadband beam steering antenna

Description:
FIELD OF THE INVENTION 
   The present invention relates to an antenna apparatus with steerable beam pattern, an RF transceiver comprising the antenna apparatus and a mobile device comprising the antenna apparatus. 
   DESCRIPTION OF THE RELATED PRIOR ART 
   The American Federal Communications Commission (FCC) allows unlicensed use of the 3.1 GHz to 10.6 GHz frequency band for ultra-wideband (UWB) applications, whereby UWB refers to a broadband radio technology having a bandwidth larger than 500 MHz or larger than 25% of the center frequency. An ultra-wideband frequency range, for example, is a frequency range having a bandwidth larger than 500 MHz or larger than 25% of the center frequency. Other nations and organizations have followed and or are expected to follow the FCC regulations. The IEEE 802.15 working group develops standards for wireless short distance or wireless personal area networks. The group&#39;s WPAN™ technology employs the 3.1 GHz to 10.6 GHz range and addresses wireless networking of portable and mobile computing devices such as PCs, PDAs, peripherals, cell phones, pagers and consumer electronics, allowing those devices to communicate and interoperate with each other and employing the 3.1 GHz to 10.6 GHz range. 
   UWB technology was at first developed in connection with radar applications. Today, however, UWB systems are also used as a wireless RF interface (e.g. wireless USB) between mobile terminals (e.g. cell phones, laptops, PDAs, wireless cameras, MP3 players) with much higher data rates than Bluetooth or IEEE 802.11. A UWB system can further be used as an integrated system for automotive in-car services, for example, as an entertainment system or any location-based system (e.g. for downloading audio or video data for passenger entertainment). 
   Traditionally, mobile and wireless handsets are equipped with a single narrowband 3D monopole or planar antenna. Planar ultra-wideband antennas including dipole, patch and bow-tie antennas and other types of planar structures are employed in a wide variety of applications today. Phased arrays that are operated with variable phase shifters are known to provide beam steering property. However, phased array antennas are relatively large in size and their integration in mobile devices (e.g. consumer electronic devices) is very challenging. 
   In view of the explanations provided above, it is the object of the present invention to provide a mobile device with a beam steerable antenna and a beam steerable antenna and RF transceiver suitable for employment in a mobile device. 
   SUMMARY OF THE INVENTION 
   The antenna apparatus according to the present invention is attachable to the front-end of a transceiver circuitry and comprises at least two balanced radiation elements forming a planar structure, for transmitting and/or receiving a corresponding number of partial signals, a signal splitter and/or combiner for splitting a signal received from an attached transceiver circuitry into said partial signals and/or combining said partial signals into a signal to be transmitted to an attached transceiver circuitry, a phase shifter device operable to apply relative phase shifts between at least two of said partial signals, whereby said relative phase shifts are selectable from a group of at least two relative phase shift values provided by said phase shifter device. 
   By providing a plurality of balanced radiation elements, a high antenna gain is provided. By providing a phase shifter device operable to apply the relative phase shifts, a plurality of radiation patterns (radiation beams) with different orientations are obtained, thus a beam steering antenna is provided. A high gain beam steering antenna reduces the power and energy needed, to operate an RF transmitter and/or receiver, thus, battery size of a mobile device can be reduced. Such antenna typically achieves a better reception in dead spots and is useful employed, for example, near walls (e.g. in a closed room) to achieve better signal reception and emission. By providing radiation elements in a planar structure, the antenna apparatus is small and is suitable for integration into mobile devices. 
   The RF transceiver according to the present invention comprises a transceiver front-end circuitry and an antenna apparatus according to the present invention wherein the transceiver front-end circuitry and the antenna apparatus are provided on a single printed circuit board. The inventive RF transceiver has, in addition to the advantages of the inventive antenna apparatus, the benefits of low cost of production, small size and high mechanical resistance (e.g. to shocks). 
   The mobile device according to the present invention comprises the antenna apparatus according to the present invention or the RF transceiver according to the present invention. 
   Advantageously comprises said signal splitter and/or combiner a Wilkinson power splitter. 
   Advantageously is said phase shifter device a broadband phase shifting device, operable in an ultra-wideband frequency range. 
   Advantageously comprises said phase shifter device a Schiffmann phase shifter. 
   Advantageously is the number of balanced radiation elements four. 
   Advantageously are the balanced radiation elements arranged in a rectangular grid. 
   Advantageously is said phase shifter device operable to apply six different nonzero phase shift values between any two of said partial signals, whereby for every one of the six different phase shift values there is another one of the six different phase shift values having the same absolute value but the opposite sign. 
   Advantageously comprises the phase shifter device a number of phase shifter banks according to the number of radiation elements, each phase shifter bank thereby comprising a plurality of selectable delay lines and operable to shift a corresponding one of said partial signals in phase by means of a selected one of said plurality of selectable delay lines. 
   Advantageously are the phase shifter banks identical. 
   Advantageously comprises each of said phase shifter banks exactly five selectable delay lines. 
   Advantageously comprises at least one of the radiation elements at least one balance element having a signal feeding point of which the width varies with the distance from the signal feeding point. 
   Advantageously are the balanced radiation elements identical. 
   Advantageously is the signal path of two partial signals between which no relative phase shift is applied mirror symmetric or point symmetric. 
   Advantageously are the balanced radiation elements adapted to emit and/or receive a radiation beam which has a vertical polarization. 
   Advantageously has a radiation beam emitted from and/or received by the balanced radiation elements a variation of the amplitude response of equal or less than 2 dBi over an ultra-wideband frequency range. 
   Advantageously has a radiation beam emitted from and/or received by the balanced radiation elements a phase variation which is linear in frequency over an ultra-wideband frequency range. 
   Advantageously provides the antenna apparatus a return of loss which is less than −10 dB in an ultra-wideband frequency range. 
   Advantageously comprises the antenna apparatus a planar reflector element parallel to the balanced radiation elements. 
   Advantageously is the reflector element located between the radiation elements and the phase shifter device and/or is the reflector element located between the balanced radiation elements and the signal splitter and/or combiner. 
   In the inventive RF transceiver, the antenna apparatus and the transceiver front-end circuitry advantageously share the core substrate of conducting material of the printed circuit board. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is explained with reference to figures of which 
       FIG. 1  shows a first embodiment of an antenna apparatus according to the present invention and an RF transceiver according to the present invention, 
       FIG. 2  shows a power splitter employed in the first embodiment, 
       FIG. 3  shows a balanced radiation element employed in the first embodiment, 
       FIG. 4  shows an antenna array with a reflector element employed in the first embodiment, 
       FIG. 5  shows a schematic of a Wilkinson power splitter employed in the first embodiment, 
       FIG. 6  shows a diagram of the phase shifts produced by coupled microstrip line and a uniform microstrip line versus the electrical length, 
       FIG. 7  shows a schematic of a phase shifter bank employed in the first embodiment, 
       FIGS. 8   a - 8   g  show 3D surface plots of the beam pattern steered in various directions, 
       FIG. 9  show the principle of arrangement of components of a second embodiment of the present invention, whereby like numbers refer to like elements in the drawings. 
   

   DESCRIPTION OF THE DETAILED EMBODIMENTS 
     FIG. 1  shows a block diagram of a first embodiment of an antenna apparatus  1  according to the present invention. The embodiment provides an ultra-wideband, high gain, directional beam steering antenna in the microwave spectrum. In this embodiment four radiation elements  10 - 1 ,  10 - 2 ,  10 - 3 ,  10 - 4  forming an array  24  of antennas are provided, however, two or more radiation elements are sufficient to implement the present invention. The antenna apparatus  1  receives and transmits an RF signal from and to the front-end of a transceiver circuitry  80 . The embodiment described is designed for a center frequency f 0  of the RF signal of 4 GHz and a bandwidth of 2 GHz. The present invention can, however, be profitably employed for frequency ranges other than 3 to 5 GHz and, especially, is not limited to the above mentioned regulatory frequency range of 3.1 to 10.6 GHz. In order to operate in a higher frequency band the antenna apparatus  1  has to be downsized and in order to operate in a lower frequency band the antenna apparatus  1  has to be upsized, as is known to the person skilled in the art (wavelength inversely proportional to frequency). The received signal is split (divided) in a power splitter  38  (not shown explicitly in  FIG. 1 , since composed of power splitters  40 - 1 ,  40 - 2 ,  40 - 3 , see  FIG. 2 ) into equal power and equal phase split signals. The present invention may, however, also be implemented with non-equal-power and non-equal-phase power splitters  38 . Each of the split signals is applied to a separate output port of the power splitter  38 , each output port connected to a separate “branch” of electronic circuitry comprising exactly one radiation element  10  of the array  24 . If a power splitter  38  does not provide equal phase split signals this can be compensated, for example, by properly designed phase shifter banks or by properly designed transmission lines. It is to be noted however, that equal phase is not necessary to implement the present invention. In case of the present embodiment, the received signal is split into four signals according to the four radiation elements  10  provided by the antenna apparatus  1 . In case of the present embodiment, the power splitter  38  is realized by three cascaded power splitters  40 - 1 ,  40 - 2 ,  40 - 3 . Each one of the power splitters  40  has three ports: one input port (P 1 ) and two output ports (P 2 , P 3 ). Besides splitting a signal that is received at the input port equally to the output ports, each one of the power splitters  40  combines (adds) signals received at the two output ports and applies the combined signal to the input port. The two output ports of the first stage power splitter  40 - 1  are connected to the two input ports of the second stage power splitters  40 - 2 ,  40 - 3 . In case of the present embodiment, the power splitters  40  are Wilkinson power splitters. Wilkinson power splitters offer the advantage of the output ports being simultaneously isolated and matched (at a given design frequency, e.g. f 0 =4 GHz). The cascaded Wilkinson power splitter offers a 6 dB loss at the end of each branch. Instead of three cascaded 3-port (2-branch) Wilkinson power splitters, a single 5-port (4-branch) Wilkinson power splitter can be employed. The power splitter  38  is formed by conductive traces (striplines/microstrips) of well-defined form and material on or in a PCB. The operational bandwidth may be increased by optimizing the conductive traces. 
   In this embodiment all branches are the same and it is understood, that if a description relating to only one branch or any element of only one branch is given, the description applies to all other branches as well. 
   The direction of maximum emission and reception of RF radiation (i.e. the direction of the radiation beam) of the antenna apparatus  1  is controlled by applying phase shifts to the signals in each branch. To this end, the embodiment provides four phase shifter banks  42 - 1 ,  42 - 2 ,  42 - 3 ,  42 - 4  according to the number of radiation elements  10  in the array  24 . In the embodiment, the phase shifter banks  42  are the same in terms of functionality provided and have essentially the same construction. The present invention may, however, also be implemented with phase shifter banks  42  which have different constructions and provide different functionality/phase shifts. In the embodiment, each phase shifter bank  42  comprises five delay lines  36 - 1 ,  36 - 2 ,  36 - 3 ,  36 - 4 ,  36 - 5  (not shown in  FIG. 1 ), which correspond to five different phase shift characteristics (phase shift dependent on frequency) which are alternatively applicable to a branch signal. If a different delay line  36  is selected in any two branches, then the signals in the respective two branches will exhibit a relative phase shift given by the difference of phase shift characteristics of the selected delay lines  36 . By this means 90°, 135° and 225° relative phase shifts are realized. 0° relative phase shifts are realized by selecting the same delay line  36  in any two branches. In each branch, power splitter side switches  44 - 1 ,  44 - 2 ,  44 - 3 ,  44 - 4  and antenna side switches  46 - 1 ,  46 - 2 ,  46 - 3 ,  46 - 4  insert one delay line  36  at a time into the signal path from the radiation element  10  to the power splitter  40 . If a delay line  36  is not inserted into the signal path, it is disconnected from the signal path at the antenna side and at the power splitter side by the antenna side switches  46  and the power splitter side switches  44 , respectively. The switches  44 ,  46  are RF switches specifically adapted to switch and transmit the RF signals of the frequency range in question. The switches  44 ,  46  are electrically controlled by an antenna controlling unit (not shown), thereby the beam steering is automated. The antenna controlling unit may be programmed to control the switches so as to scan all possible directions and lock to the direction with the best received signal strength. The phase shifter banks  42  (i.e. the delay lines  36 ) are formed by conductive traces (striplines/microstrips) of well-defined form and material on or in a PCB. In the embodiment, each phase shifter bank  42  provides five different phase shift characteristics. The present invention may, however, also be implemented with two or more different phase shift characteristics. Also, some branches may be provided with a phase shifter bank while others may not. 
   The signal received from and transmitted to the transceiver circuitry is an unbalanced signal, the radiation elements  10  are of the dipole type and operate with a balanced signal, therefore a conversion is performed. The branch signals are feed to and collected from the radiation elements  10  by means of unbalanced-balanced microstrips  48 - 1 ,  48 - 2 ,  48 - 3 ,  48 - 4 . These microstrips  48  provide a conversion from an unbalanced signal to a balanced signal and vice versa. Other balun-type devices may be employed however. 
   In the embodiment, a reflector element  26  (not shown in  FIG. 1 ) provided in proximity of the antenna array  24 . The reflector element  26  partly shields the radiation elements  10  and modifies the directional characteristic and frequency response of the antenna array  24 . The reflector element  26  may be at floating potential or may be connected to ground potential. 
   The embodiment provides a symmetric arrangement.  FIG. 1  shows an X- and a Y-axis of an orthogonal coordinate system further comprising a Z-axis (orthogonal to the drawing plane) corresponding to—as a manner of speaking—a “height”. The power splitters  40 , the switches  44 , the switches  46 , the balanced to unbalanced microstrips  48 , the radiation elements  10 , the reflector element  26  and the transmission lines (including the elements in these components, e.g. the delay lines  36 ) each are arranged mirror symmetric with respect to a Y-plane (Y=0) comprising the X-axis and the Z-axis and/or are arranged mirror symmetric with respect to an X-plane (X=0) comprising the Y-axis and the Z-axis and/or are arranged point symmetric within the Z-plane (Z=0) with respect to the origin (X=0, Y=0). Which components obey which symmetry can be derived from  FIG. 1  and  FIG. 4 . For example, the corresponding components in the first branch and the fourth branch (e.g. the phase shifter banks  42 - 1  and  42 - 4 ) are arranged mirror symmetric with respect to the X-plane. As another example, the corresponding components in the first branch and in the second branch (e.g. the switches  44 - 1  and  44 - 2 ) are arranged mirror symmetric with respect to the Y-plane. As still another example, the corresponding components of the first branch and the third branch (e.g. the transmission lines between the components) are arranged point symmetric. As a last example, the power splitters  40 - 2  and  40 - 3  are arranged mirror symmetric with respect to the X-plane and point symmetric. Thus, the signal path of two branch signals to which no relative phase shift is applied is symmetric (mirror and/or point) in space. Therefore, the time needed for design and testing of the antenna apparatus  1  decreases und, thus, the price of the antenna apparatus  1  is reduced. Because of the symmetry of the radiation elements  10 , the main beam pattern (see below) exhibits symmetry and the set of possible beam pattern directions exhibit symmetry. 
   In the embodiment, the power splitter  38 , the phase shifter banks  42 , the antenna feeds  48 , the radiation elements  10 , the reflector element  26  and the transmission lines connecting these elements are formed by conductive traces (striplines/microstrips) of well-defined form and material on or in a single PCB. Therefore, the present invention can be cheaply manufactured, is highly integrated and small (especially flat) and highly resistant to shocks and other mechanical wear. By using a common layout procedure and a common substrate, the antenna print and the classical RF front-end circuitry  80  can be simultaneously manufactured, so that a substantial cost reduction is achieved. 
   Alternatively, a separate antenna module comprising the radiation elements  10  and the microstrips  48  and, eventually, the reflector element  26  may be provided. In this case, the microstrips  48  may be connected to the feeding network (i.e. the switches  44 ,  46 , the phase shifter banks  42 , the power splitter  38  and the interconnections) by a coaxial cable or a mini-SMP connector. 
     FIG. 3  shows a balanced radiation element (dipole type antenna)  10  consisting of two conducting balance elements  12 ,  14 . The balanced radiation element  10  is described with the help of an Y′-Y′-Z′ orthogonal coordinate system which differs from the X-Y-Z coordinate system only by a translation. The balanced radiation element  10  is essentially flat and is confined within a small region around the Z-plane (Z=0). The balanced radiation element  10  is mirror symmetric with respect to the Y′-axis which extends along the length of the balanced radiation element  10 . Thereby, each of the balance elements  12 ,  14  is mirror symmetric with respect to the Y′-axis. The balanced radiation element  10  is mirror symmetric with respect to the X′-axis which extends along the width of the balanced radiation element  10 . Thereby, one of the balance elements  12 ,  14  is a mirror image of the other one of the balance elements  12 ,  14 . Both balance elements  12 ,  14  may, for example, be formed on one side of a (planar) printed circuit board (PCB). Alternatively, balance element  12  may be formed on the bottom surface of a PCB and balance element  14  may be formed on the top surface of a PCB or vice versa. In the latter case, the thickness of the PCB should be small compared to a characteristic dimension of the radiation element  10  as will be readily acknowledged by the skilled person. In the latter case still, the radiation element  10  point symmetrical with respect to the origin of the X′-Y′-Z′ coordinate system, so that the balance element  14  is the point symmetrical image of the balance element  12 . In both cases, the balance element  12  and the balance element  14  have the same shape and each of the balance elements  12 ,  14  is mirror symmetric with respect to an axis along the length of the balanced radiation element. 
   The balance elements  12 ,  14  have essentially the same shape and are made from the same material(s), for example, copper, aluminium and/or other metallic components. Thus, in the following, the balance element  12  is described and the description of balance element  14  is omitted and it is understood that the description of balance element  12  applies to balance element  14  where applicable. The balance element  12  is essentially flat. The balance element  12  has an inner or center end  16 . The balance element  12  is feed at or near the center end  16  with an electric signal by a microstrip feed line (not shown) which is connected to the balance element  12  at or near to the center end  16 . The inner end  16  of the balance element  12  is opposing the corresponding inner end of the balance element  14 . The balance element  12  has an outer end  18 , which is opposing the inner end  16 . The balance element is tapering from the outer end  18  to the inner end  16  in order to achieve broadband impedance matching and provide a large bandwidth antenna. Thus, the width of the balance element  12  is higher at the outer end  18  than at the inner end  16 . In the embodiment described, the balance element  12  has the specific shape of a triangle  20  of which one corner (the inner end corner) is cut away and replaced by a rectangle  22 . The rectangle portion  22  is flush with the (cut) triangle portion  20 . Thus, the shape of balanced radiation element  10  of the embodiment is resembling a bow tie. However, the present invention is not limited to bow type antennas. Another example, is a balanced antenna radiator formed by two rhombi, arranged such that the corresponding diagonals of the rhombi are aligned along the length, whereby the rhombi are feed at the inner, opposing corners. However, bow type antenna has the advantage of being shorter in length and, thus, providing a smaller size of the antenna apparatus. 
     FIG. 4  shows an array  24  of antennas and a reflector element  26 . The array  24  comprises four balanced radiation elements  10 - 1 ,  10 - 2 ,  10 - 3 ,  10 - 4 . The four balanced radiation elements are identical among themselves and are identical to the balanced radiation element  10  described above. Therefore, if not a specific one of the balanced radiation elements is desired to be addressed, it is simply referred to balanced radiation element  10  and the set of the balanced radiation elements is simply referred to as balanced radiation elements  10  (the same convention is adopted for the power splitters  40 , the phase shifter banks  42 , the power splitter side switches  44 , the antenna side switches  46  and the balanced to unbalanced microstrips  48 ). The orientation of each of the balanced radiation elements  10  is the same as in  FIG. 3 . That is, the length of each of the balanced radiation elements  10  is along the Y-axis and the width of each of the balanced radiation elements  10  is along the X-axis. Also, the balanced radiation elements  10  are located at the same height at Z=0. Thus, the antenna array  24  is a planar device like the balanced radiation elements  10  and can be easily fabricated on a PCB, for example, by etching copper on a dielectrical substrate. 
   The balanced radiation elements  10  are arranged in a rectangular grid. The grid length in X-direction is greater than the width of the balanced radiation element  10  and the grid length in Y-direction is greater than the length of the balanced radiation element  10 . The distance between the radiation elements  10  is optimized to achieve high gain and impedance matching in the whole frequency band. A grid length of (0.63+/−0.3)*λ 0  in X-direction and (0.70+/−0.3)*λ 0  in Y-direction has been shown to be advantageous, whereby λ 0  is the wavelength at the center frequency f 0  (e.g. 4.7 cm and 5.2 cm at f 0 =4 GHz). 
   Located below and spaced from the balanced radiation elements  10  by a distance h&gt;0 is the reflector element  26 . The reflector element  26  may be made from any conducting material, including, for example, copper, aluminium and/or other metallic components. Preferably, the reflector element  26  is essentially flat and parallel to the X-Y-plane, that is, the reflector element  26  is preferably parallel to the plane in which the antenna array  24  lies. Preferably, the reflector element  26  extends at least just beyond the balanced radiation elements  10 , has no holes and/or is of a convex shape. The planar reflector element  26  acts as a mirror to RF waves and reflects the radiation pattern in one plane, thus, assists in providing a high antenna gain. A high value of the reflector element&#39;s  26  surface impedance to electromagnetic waves is advantageous. The reflector plane  26  may extend considerably beyond the balanced radiation elements  10 . 
   The reflector element  26  may for example have a rectangular shape as depicted in  FIG. 4 . The reflector element  26  may, for example by formed by etching copper on a dielectric substrate. The distance h is optimized in order to meet the specifications. 
   This type of antenna is able to achieve a bandwidth of more than 50% of the center frequency f 0  at a voltage standing wave ratio (VSWR) of 2:1. For a higher bandwidth, the impedance matching can be improved by modifying the shape of the radiation elements  10 , for example, by smoothing the angles of the radiation elements  10 . 
   The balanced radiation element  10  is feed by a balanced to unbalanced microstrip  30 . The balanced to unbalanced microstrip  30  comprises a first conductor connected to the first balance element  12  and a second conductor connected to the second balance (element  14 . The first and second conductors run parallel and close to each other. At one end, the first and second conductors are connected to or near to the inner ends  16  of the balance elements  12 ,  14 . The first and second conductors are orthogonal to the length of the balanced radiation element  10 . In case that the balance elements  12 ,  14  are located the top and the bottom side of a PCB, the first and the second conductors may too be located on the top and on the bottom side of the PCB, respectively. The construction and the application of a balanced to unbalanced microstrip  30  are known to the skilled person. A further description thereof is therefore omitted. 
     FIG. 5  shows a schematic diagram of one of the cascaded Wilkinson power splitters  40 , which applies to each of the three cascaded Wilkinson power splitters  40 . In the Wilkinson power splitter  40 , the input port (P 1 ) and the first output port (P 2 ) are connected by a first microstrip line  32 - 1 , the input port and the second output port (P 3 ) are connected with a second microstrip line  32 - 2  and the first output port and the second output port are connected by a resistor  34  also formed by a microstrip line. The first and the second microstrip lines  32  are quarter wave transformers (i.e. apply a 90° phase shift) with a characteristic impedance of √{square root over (2)}*Z 0  and the resistance of the resistor  34  is 2*Z 0 , whereby Z 0  is the characteristic impedance of the power splitter  40 . Impedance matching is achieved, when all ports of the power splitter are terminated with a characteristic impedance of Z 0 . It is to be noted, that the advantageous properties of the Wilkinson Power splitter of the output ports being isolated and matched are strictly valid only at a given design frequency (e.g. f 0 =4 GHz) (the more the frequency is distinct from the design frequency, the more the properties are violated). Refinements of the basic design of  FIG. 3  are known which provide for a more broadband Wilkinson power splitter than the principle design of  FIG. 3 . However, the basic design has been shown to be perform sufficiently well to obtain an ultra-wideband antenna apparatus ( 1 ). 
   The generation of the relative phase shifts of 90°, 135° and 225° is explained with reference to  FIGS. 6 and 7 . 
   The type of phase shifter used are called Schiffman phase shifters (IRE Trans. MTT April 1958). These phase shifters employ a section of coupled microstrip transmission lines as key elements. The coupled lines of a Schiffman phase shifter are parallel, have equal length l and are connected at one end. The other end is used as input and output of the network (coupled lines seen as network). Since connected at one end, the two coupled lines may simply be called a coupled line. The image impedance Z 1  and the phase shift φ of such a coupled line is given by 
             Z   I     =         Z     0   ⁢           ⁢   o       ⁢     Z     0   ⁢           ⁢   e                     and                 cos   ⁢           ⁢   ϕ     =           Z     0   ⁢           ⁢   e         Z       0   ⁢           ⁢   o     ⁢                 -       tan   2     ⁢     θ   el               Z     0   ⁢           ⁢   e         Z     0   ⁢           ⁢   o         +       tan   2     ⁢     θ   el             ,         
whereby Z 0o  and Z 0e  are the odd and even characteristic impedances of the coupled line, θ el =β*l is the electrical length of each of the coupled lines and β is the phase constant. This differs from a uniform microstrip line, which produces a phase shift that is proportional to the electrical length.  FIG. 6  shows a plot of the phase shifts  35  produced by a coupled line and of a uniform line versus the electrical length θ el . It can be seen that there is a large range (approx. 45° to 135°) in the electrical length where the phase characteristic  35 - 1  of the coupled line is approximately parallel to the phase characteristic  35 - 2  of the uniform microstrip line. In this range, the phase difference is approximately constant. As the phase constant is proportional to the frequency of a signal, a constant phase shift is obtained for a large frequency bandwidth (here: 100% of center frequency). The same principle can be applied to two coupled line networks with a given length.
 
     FIG. 7  shows a schematic of the phase shifter bank  42  of the embodiment of the present invention. The phase shifter bank  42  comprises three coupled microstrip lines  36 - 1 ,  36 - 2 ,  36 - 3  and two uniform microstrip lines  36 - 4 ,  36 - 5 , which, together, form the five delay lines  36 . The first coupled line  36 - 1  and the first microstrip line  36 - 4  are used to generate the 225° relative phase shift, the second coupled line  36 - 2  and the second microstrip line  36 - 5  are used to generate the 135° relative phase shift and the third coupled line  36 - 3  and the second microstrip line  36 - 5  are used to generate the 90° relative phase shift. Thus, the second microstrip line  36 - 5  serves the generation of the 90° and 135° relative phase shifts. Alternatively, separate uniform microstrip lines could be provided for the generation of the 90° and 135° phase shifts. In this alternate case, there are six delay lines  36  in total with three coupled microstrip lines and three corresponding uniform microstrip lines. However, having the microstrip line  36 - 5  serve a double purpose saves space and reduces the amount of paths to be switched, thus, simplifies the RF switches  44 ,  46 . In order to apply a phase shift between any two of the radiation elements  10 , the coupled line corresponding to the desired phase shift is inserted into the signal path to/from one of the two radiation elements and the uniform microstrip line corresponding to the desired phase shift is inserted into the signal path to/from the other of the two radiation elements. For example, if a 90° phase shift is to be applied between the radiation elements  10 - 1  and  10 - 4 , the switches  44 - 1  and  46 - 1  insert the coupled line  36 - 3  into the first branch (to/from radiation element  10 - 1 ) and the switches  44 - 4  and  46 - 4  insert the microstrip line  36 - 5  into the fourth branch (to/from radiation element  10 - 4 ). In order to obtain the reverse shift of −90°, the switches  44 - 1  and  46 - 1  insert the microstrip line  36 - 5  into the first branch (to/from radiation element  10 - 1 ) and the switches  44 - 4  and  46 - 4  insert the coupled line  36 - 3  into the fourth branch (to/from radiation element  10 - 4 ). It can be seen, that although each phase shifter bank  42  provides the essential elements of a Schiffman phase shifters (e.g. the coupled line  36 - 1  and the microstrip line  36 - 4  may be seen as forming a 225° Schiffman phase shifter), the Schiffman phase shifters as employed in this embodiment are not located within a single phase shifter bank, but are dispersed over the phase shifter banks  42 . 
   The described embodiment of the present invention is operable to electronically steer the beam pattern in 7 different directions by varying the phase shift characteristic applied to the signal in each branch (only the relative phase of the branch signals is relevant). For all directions, the beam width is approximately 40°. The orientation of the beam pattern is described with reference to  FIGS. 8   a  to  8   g . For this purpose the coordinate system with axes X, Y and Z defined above is described in spherical coordinates, whereby the X-Y plane forms a horizontal plane and corresponds to an angle of elevation θ=0° and the positive X-axis direction corresponds to an azimuth angle φ=0°. 
     FIG. 8   a  shows the orientation of the main beam (θ=90°). The direction of maximum emission/reception of the main beam is orthogonal to the plane of the antenna array  24 , orthogonal to the reflector plane  26  and points away from the reflector element  26 . The main beam is obtained by selecting the same phase shifter characteristic (the same delay line  36 ) for all radiation elements  10 . 
   When a +/−90° phase shift is applied between radiation elements  10 - 1  and  10 - 2  and between the radiation elements  10 - 4  and  10 - 3 , the beam pattern is tilted by approximately 30° from the main beam at azimuth angles of 0° and 180°. (θ=60°, φ=0°, 180°). This is shown in  FIG. 8   b  and  FIG. 8   c.    
   When a phase shift of +/−135° is applied between the radiation elements  10 - 1  and  10 - 2  and a phase shift of +/−90° is applied between the radiation elements  10 - 4  and  10 - 3 , the beam pattern is tilted by approximately 30° from the main beam at azimuth angles of approximately 40° and 320° (θ=60°, φ=40°, 320°). This is shown in  FIGS. 8   d  and  8   e.    
   When a phase shift of +/−90° is applied between the radiation elements  10 - 1  and  10 - 2  (and a phase shift of +/−225° is applied between the radiation elements  10 - 4  and  10 - 3  the beam pattern is tilted by approximately 30° from the main beam at azimuth angles of approximately 140° and 220° (θ=60°, φ=140°, 220°). This is shown in  FIGS. 6   f  and  6   g.    
   The embodiment provides a beam steering directional radiation pattern in azimuth plane with 360° in elevation over the entire frequency range. The radiation beam thereby exhibits linear polarization and a linear phase variation Δφversus frequency ω, thus, a constant group delay 
                     τ   g     ⁡     (   ω   )       =         ⅆ     φ   ⁡     (   ω   )           ⅆ   ω       =         τ     g   ⁢           ⁢   0       ⁢           ⁢   with   ⁢           ⁢     τ     g   ⁢           ⁢   0         =     const   .                 (   1   )               
over the entire frequency range, as well as a flat amplitude response over the entire frequency range (the antenna gain ranges from 6 to 8 dBi, i.e. the variation of the amplitude response is not more than 2 dB at the direction of maximum emission/reception). Without using a resistive loading, the return loss
   RL=− 20·log 10 | ρ | [dB],  (2a) 
which is defined over the magnitude of the complex-valued reflection coefficient  ρ  as the ratio (in dB) of the power incident on the antenna terminal to the power reflected from the antenna terminal, has a value of less than −10 dB in a frequency range between 3 and 5 GHz, which corresponds to a voltage standing wave ratio
 
                 VSWR   =       1   +          ρ   _              1   -          ρ   _                      (     2   ⁢           ⁢   b     )               
of less than 2.
 
   The embodiment fulfills the FCC regulations and the IEEE 802.15 WPAN standards for the 3 to 5 GHz frequency range. The embodiment further provides a high antenna efficiency and allows for the control of the specific absorption rate (SAR) so that compliance with the FCC standards on mobile headset emission is easily achieved for devices equipped with it. 
   In a second embodiment, the antenna apparatus ( 2 ) is provided with a sandwiched structure as shown in  FIG. 9 . Here, at least part of the antenna feeding network  50  (i.e. the switches  44 ,  46 , the phase shifter banks  42 , the power splitter  38  and the interconnections) is located below the reflector element  26 , thus a layered structure with the reflector element  26  in between the radiating elements  10 - 1 ,  10 - 2 ,  10 - 3 ,  10 - 4  and the feeding circuitry is obtained, which reduces the area needed for the antenna apparatus. 
   This layered structure can be integrated by filling the spaces between the network  50 , the reflector plane  26  and the radiating elements  10  with electrically non-conducting material (insulator, semiconductor, . . . ). Thus the layered structure can be provided as a layered board structure. 
   The connection of the radiating elements  10  to the feeding circuitry may be around the reflector element  26  or by piercing the reflector element  26 . Besides of this layer structure and any difference that might arise as a logical consequence of the layer structure, the second embodiment is the same as the first embodiment. Especially, the corresponding components in each branch in the second embodiment are arranged in a symmetrical manner as in the first embodiment. 
   The antenna apparatus of the present invention can be advantageously employed in any mobile computing or communication devices such as, for example, PCs, PDAs, peripherals, cell phones, pagers and consumer electronics for providing a wireless RF interface. However, the antenna apparatus may also be advantageously employed in non-mobile devices. 
   The present invention has been explained with reference to specific embodiments, this is by way of illustration only and it will be readily apparent to those skilled in the art that various modifications may be made therein without departing from the scope of the following claims.