Patent Publication Number: US-2023136027-A1

Title: Protecting Multi-Level Power Converters

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS - CLAIM OF PRIORITY 
     The present application claims priority to U.S. provisional Pat. Application No. 63/274,251, filed on Nov. 1, 2021, for a “Protecting Multi-Level Power Converters”, which is herein incorporated by reference in its entirety. 
    
    
     BACKGROUND 
     Technical Field 
     This invention relates to electronic circuits, and more particularly to power converter circuits, including DC-DC power converter circuits. 
     Background 
     Many electronic products, particularly mobile computing and/or communication products and components (e.g., notebook computers, ultra-book computers, tablet devices, LCD and LED displays) require multiple voltage levels. For example, radio frequency (RF) transmitter power amplifiers may require relatively high voltages (e.g., 12 V or more), whereas logic circuitry may require a low voltage level (e.g., 1-3 V). Still other circuitry may require an intermediate voltage level (e.g., 5-10 V). 
     Direct current power converters are often used to generate a lower or higher voltage from a common power source, such as a battery, solar cells, and rectified AC sources. Power converters which generate a lower output voltage level from a higher input voltage power source are commonly known as buck converters, so-called because the output voltage V OUT  is less than the input voltage V IN , and hence the converter is “bucking” the input voltage. Power converters which generate a higher output voltage level from a lower input voltage power source are commonly known as boost converters, because V OUT  is greater than V IN . Some power converters may be either a buck converter or a boost converter depending on which terminals are used for input and output. Some power converters may provide an inverted output. 
     Various types of power converter comprise a converter circuit, control circuitry, and auxiliary circuitry such as bias voltage generator(s), a clock generator, a voltage regulator, a voltage control circuit, etc. For example,  FIG.  1    is a block diagram of a prior art power converter  100 . A converter circuit  102  having input terminals T1/T1′ and output terminals T2/T2′ is configured such that input terminals T1/T1′ are coupled to a voltage source  104  (e.g., a battery) and output terminals T2/T2′ are coupled to an output capacitor C OUT  and a load  106  (which also may be represented as an equivalent resistance R). 
     In the illustrated example, the voltage applied across input terminals T1/T1′ is V IN , and the converted voltage provided across output terminals T2/T2′ is V OUT . A controller  108  outputs a set of control-signals  112  that control the internal components of the converter circuit  102  (e.g., internal switches, such as FETs, especially MOSFETs) to cause the converter circuit  102  to either boost or buck V IN  to V OUT . The controller  108  may also receive a separate set of control signals  112  indicative of the converter circuit  102  operational status. An auxiliary circuit  110  may provide various voltages and/or signals to the controller  108  (and optionally directly to the converter circuit  102 ), such as a voltage V DD , a clock signal CLK, a circuit bias voltage V BIAS , and one or more control signals CTRL. Power to the illustrated auxiliary circuit  110  is supplied at terminal V INPUT , and may come from the illustrated voltage source  104  or from another source (including the converter circuit  102  when fully operational). 
     The converter circuit  102  may be, for example, a switched-capacitor network such as a charge pump, or an inductor-based regulator. Multi-level converters typically combine a charge-pump capacitive voltage converter with an inductor-based power converter in one structure, allowing lower voltage power switches to be used. In many designs, the converter circuit  102  uses capacitors as energy storage elements to transfer charge from the input to the output of the power converter  100 . These charge transfer capacitors are commonly known as “fly capacitors” or “pump capacitors” and may be external components coupled to an integrated circuit embodiment of the power converter  100 . In some designs, the converter circuit  102  also uses an inductor as an energy storage element. 
     A general problem with many FET-based power converter architectures is that excessive current in-rush needs to be avoided during startup of the power converter. In integrated circuit implementations, parasitic inductances exist (for example, due to on-die conductor routing and printed circuit board conductor routing). While the parasitic inductances increase the total impedance in the circuit and thus help reduce the maximum current, the parasitic inductances also store energy which extends how long a current surge is present and can result in ringing if there is capacitance present (which is the case in such circuits). Such ringing can cause voltage spikes that are significantly higher than V IN  and electrically overstress the charge pump power switches, affecting their reliability, potentially to destruction. In addition, voltage spikes higher than V IN  due to rapidly changing current flow may occur in other cases, such as when the electrical charge among the fly capacitors is out of balance or when the output of the converter circuit is shorted. 
     A conventional solution to the problem of voltage spikes is to require that all of the FET power switches in a multilevel power converter be high-voltage devices (e.g., having a breakdown voltage BV DS  of equal to or higher than the maximum designed V IN ). A drawback of such high-voltage devices is that they consume more die area - which generally means higher cost -and are generally less efficient than lower voltage breakdown FETs. 
     It would be desirable protect a multi-level power converter using no more than two high-voltage FET switches while allowing all or most other switches to be low-voltage FET switches. The present invention addresses this need. 
     SUMMARY 
     The present invention encompasses circuits and methods for protecting a multi-level power converter using no more than two high-voltage FET switches while allowing all or most other power switches to be low-voltage FET switches. Some embodiments provide protective high-voltage top and bottom FETs designed to saturate before the remaining low-power FET switches Swx saturate. Other embodiments may use only low-power FETs for the power switches Swx but provide protective circuits configured to be in an always-ON (conducting) state when in normal power conversion operation, and to quickly switch to an OFF (non-conducting) state in the event of transients or a fault condition. Optionally, one or more of the protective circuits may be used in a controlled manner to limit or block current flow during certain types of fault conditions and/or to limit in-rush current during startup of a power converter. 
     Embodiments include a multi-level power converter based on field-effect transistors (FET), including: a first protective FET switch configured to be coupled to an input voltage source, the first protective FET switch having a first breakdown voltage characteristic and a first saturation current characteristic; a second protective FET switch configured to be coupled to a reference voltage (e.g., circuit ground), the second protective FET switch having a second breakdown voltage characteristic and a second saturation current characteristic; and a set of one or more intermediate series-coupled FET switches, the set being series-coupled between the first protective FET switch and the second protective FET switch, wherein one or more of the FET switches in the set have a breakdown voltage characteristic that is less than the first breakdown voltage characteristic and the second breakdown voltage characteristic, and have a saturation current characteristic that is greater than the first saturation current characteristic and the second saturation current characteristic. 
     Other embodiments include a multi-level power converter, including a first protective switch configured to be coupled to an input voltage source and having a control input for setting the first protective switch to an open state or to a closed state; a second protective switch configured to be coupled to a reference voltage (e.g., circuit ground) and having a control input for setting the first protective switch to an open state or to a closed state; and a set of one or more series-coupled field-effect transistors (FET) switches, the set being series-coupled between the first protective switch and the second protective switch; wherein the first protective switch and the second protective switch are set to the closed state during normal operation of the multi-level power converter, and at least one of the first protective switch and the second protective switch is set to the open state in response to one or more transient events or fault events. 
     Still other embodiments include a multi-level power converter based on field-effect transistors (FET), including a first FET protective switch configured to be coupled to an input voltage source and having a control input for setting the first FET protective switch to an open state or to a closed state; a second FET protective switch configured to be coupled to a reference voltage (e.g., circuit ground) and having a control input for setting the second FET protective switch to an open state or to a closed state; and a set of one or more series-coupled FET power switches, the set being series-coupled between the first FET protective switch and the second FET protective switch; wherein the first FET protective switch and the second FET protective switch are set to the closed state during normal operation of the multi-level power converter, and wherein at least one of the first FET protective switch and the second FET protective switch is set to the open state in response to one or more transient events or fault events. 
     Yet other embodiments include a multi-level power converter based on field-effect transistors (FET), including a set of one or more series-coupled FET power switches, the set including a first FET power switch configured to be coupled to an input voltage source and a last FET power switch configured to be coupled to a reference voltage (e.g., circuit ground); a first FET protective switch having a control input for setting the first FET protective switch to an open state or to a closed state, the first FET protective switch coupled between the first FET power switch and a next FET power switch in the set of one or more series-coupled FET power switches; and a second FET protective switch configured to be coupled to a reference voltage (e.g., circuit ground) and having a control input for setting the second FET protective switch to an open state or to a closed state, the second FET protective switch coupled between the last FET power switch and a preceding FET power switch in the set of one or more series-coupled FET power switches; wherein the first FET protective switch and the second FET protective switch are set to the closed state during normal operation of the multi-level power converter, and wherein at least one of the first FET protective switch and the second FET protective switch is set to the open state in response to one or more transient events or fault events. 
     The details of one or more embodiments of the invention are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the invention will be apparent from the description and drawings, and from the claims. 
    
    
     
       DESCRIPTION OF THE DRAWINGS 
         FIG.  1    is a block diagram of a prior art power converter. 
         FIG.  2    is a schematic diagram of one type of prior art multi-level converter circuit that may be used as the converter circuit of  FIG.  1   . 
         FIG.  3    is a block diagram of a first embodiment of a multi-level converter circuit with protection circuitry in accordance with the present invention. 
         FIG.  4    is a block diagram of a second embodiment of a multi-level converter circuit with protection circuitry in accordance with the present invention. 
         FIG.  5    is a block diagram of a third embodiment of a multi-level converter circuit with protection circuitry in accordance with the present invention. 
         FIG.  6    is a block diagram of a fourth embodiment of a multi-level converter circuit with protection circuitry in accordance with the present invention. 
         FIG.  7    is a block diagram of one embodiment of control circuitry for an M-level converter cell coupled to an output block comprising an inductor L and an output capacitor C OUT . 
         FIG.  8    is a process flow chart showing a first method for protecting a power converter. 
         FIG.  9    is a process flow chart showing a second method for protecting a power converter. 
         FIG.  10    is a process flow chart showing a third method for protecting a power converter. 
         FIG.  11    is a process flow chart showing a fourth method for protecting a power converter. 
     
    
    
     Like reference numbers and designations in the various drawings indicate like elements. 
     DETAILED DESCRIPTION 
     The present invention encompasses circuits and methods for protecting a multi-level power converter using no more than two high-voltage FET switches while allowing all or most other power switches to be low-voltage FET switches. Some embodiments provide protective high-voltage top and bottom FETs designed to saturate before the remaining low-power FET switches Swx saturate. Other embodiments may use only low-power FETs for the power switches Swx but provide protective circuits configured to be in an always-ON (conducting) state when in normal power conversion operation, and to quickly switch to an OFF (non-conducting) state in the event of transients or a fault condition. Optionally, one or more of the protective circuits may be used in a controlled manner to limit or block current flow during certain types of fault conditions and/or to limit in-rush current during startup of a power converter. 
     General Approach 
       FIG.  2    is a schematic diagram of one type of prior art multi-level converter circuit  200  that may be used as the converter circuit of  FIG.  1   . In the illustrated example, which is a 4-level multi-level DC-to-DC converter, the converter circuit  200  switches between multiple states, each of which defines one of several different voltages presented to an inductor L coupled to an output capacitor C OUT  and a load resistance R. The illustrated multi-level DC-to-DC converter may be configured as a buck only converter, as a boost only converter, or as a buck-boost converter, and may be inverting or non-inverting. 
     In greater detail, the multi-level converter  200  converts an input voltage V IN  to an output voltage V OUT  by actively switching two or more series-connected power switches (generally MOSFETs). The state transition patterns of the power switches determine operational zones and corresponding V OUT  ranges. In the example of  FIG.  2   , the multi-level converter circuit  200  includes an inductor L bracketed by two sets of series-connected power switches, S w   1 -S w   3  and S w   4 -S w   6  (generically, Swx). Each pair of power switches in the first and second sets of series-coupled power switches is separated by a respective node. The node between each pair of power switches in one set is coupled by a fly capacitor C FX  to the corresponding node between each pair of power switches in the other set. For example, in  FIG.  2   , node A between power switches S w   2  and S w   3  is coupled by fly capacitor C F1  to node A′ between power switches S w   4  and S w   5 ; similarly, node B between power switches S w   1  and S w   2  is coupled by fly capacitor C F2  to node B′ between power switches S w   5  and S w   6 . Controlling clock signals for each set of power switches during normal conversion operation are generally non-overlapping and complementary, and are provided by a control circuit (e.g., the controller  108  of  FIG.  1   ). 
     The power switches (such as power switches S w   1 -S w   6  in  FIG.  2   ) in modern power converters (particularly multi-level power converters) are often implemented with FETs, especially MOSFETs. It generally would be advantageous to allow the use of low-voltage FETs (e.g., having a breakdown voltage BV DS  substantially lower then V IN ) wherever possible as consuming less integrated circuit die area and generally being more efficient than higher breakdown FETs. A disadvantage of using low-voltage FETs is that they require protection from damage during stress conditions, particular from voltages that exceed the breakdown voltage of such FETs. 
     For example, as mentioned above, a general problem with many FET-based power converter architectures is that excessive current in-rush needs to be avoided during startup of the power converter. For instance, absent sufficient guard circuitry, when V IN  is first applied to a power converter of the type shown in  FIG.  1   , none of the fly capacitors C FX  would be initially charged and accordingly current may rush into the circuit. For example, if the total sum of the R ON  values of the FET power switches is 10 milliohms (0.01 ohms), and V IN  is 10 V, then as a result of Ohm’s law, V = I × R, the in-rush current will be a spike of about 1,000 amps. In integrated circuit implementations, parasitic inductances exist (for example, due to on-die conductor routing and printed circuit board conductor routing). While the parasitic inductances increase the total impedance in the circuit and thus help reduce the maximum current, the parasitic inductances also store energy which extends how long the current surge is present and can result in ringing if there is capacitance present (which is the case in such circuits). Such ringing can cause voltage spikes that are significantly higher than V IN  and electrically overstress the charge pump power switches, affecting their reliability, potentially to destruction. In addition, voltage spikes higher than V IN  due to rapidly changing current flow may occur in other cases, such as when the electrical charge among the fly capacitors C FX  is out of balance or when the output of the converter circuit is shorted. 
     In multi-level power converters like the circuit shown in  FIG.  2   , the fly capacitors C FX  at times are referenced to the V IN  supply and at times are referenced to ground. During stable operation, the FET switches bracketed by (i.e., in parallel with) the fly capacitors C FX  see less voltage across them due to the voltage stabilizing effects of the fly capacitors C FX . However, during certain transients or fault conditions, the fly capacitors C FX  may no longer hold off the voltage sufficiently to avoid damage to the FET switches. For example, during and/or after a large input supply increase (such as when changing from one power source to another power source having more or less voltage), the fly capacitors C FX  need time to the re-balance their voltages with respect to each other and with respect to C OUT ; until re-balancing occurs, low-voltage FETs may be stressed. As another example, during an output short condition, resulting high current spikes may overstress the FET switches despite the presence of the fly capacitors C FX . Moreover, the “top” and “bottom” FET switches S w   1  and S w   6  (shown encircled by dotted lines in  FIG.  2   ) have no bracketing protective fly capacitor. 
     It was realized that, in contrast to conventional solutions requiring high-voltage FETs for all power switches, low-voltage FETs could be used for most or all of the power switches Swx in a multi-level power converter if current surges induced by sudden voltage changes could be controlled, particularly in light of the voltage stabilizing effects of the fly capacitors C FX  during normal converter operation. This insight led to several solutions described below. 
     First Embodiment 
       FIG.  3    is a block diagram of a first embodiment  300  of a multi-level converter circuit with protection circuitry in accordance with the present invention. The illustrated multi-level converter circuit is a 4-level multi-level DC-to-DC converter similar in many aspects to the multi-level converter circuit  200  of  FIG.  2   . Intermediate FET power switches S w   2 -S w   5  in the illustrated example may be low-power FETs (e.g., having a breakdown voltage BV DS  as low as 1/3V IN ). However, the top power switch S w   1  and the bottom power switch (S w   6  in this example) should be capable of withstanding the worst-case voltage condition to which the power converter might reasonably be subjected in an intended application (e.g., having a breakdown voltage BV DS  greater or equal to V IN ), and optionally should be selected to saturate before any of the remaining power switches saturate (i.e., switches S w   2 -S w   5  in the illustrated example). Both protective switches S w   1 , S w   6  may be identical with respect to breakdown voltage and saturation current characteristics, but need not be identical in some applications. While protective switches S w   1 , S w   6  are shown as FETs in  FIG.  3   , in some embodiments, the protective switches S w   1 , S w   6  may be implemented in other technologies, such as bipolar junction transistors (BJTs). 
     In the illustrated embodiment, the gates (control inputs) of the top and bottom protective power FET switches are shown coupled to respective switch controllers  302   a ,  302   b , which may include suitable voltage level shifters and drivers as may be needed to open or close the associated FETs. The illustrated switch controllers  302   a ,  302   b  are shown as distinct circuit blocks, but may be implemented as a single circuit block. The switch controllers  302   a ,  302   b  are shown coupled to one or more sensing circuits  304   a ,  304   b  that may cause the switch controllers  302   a ,  302   b  to force open one or both of the top and bottom protective switches (S w   1  and S w   6  in this example) in the event of a transient or fault condition. Such action alone would provide protection for the intermediate FET power switches. However, as discussed in greater detail below, selecting high-voltage and lower saturation current characteristics for the bracketing top and bottom protective switches may alone provide protection for the intermediate FET power switches, and accordingly the switch controllers  302   a ,  302   b  and sensing circuits  304   a ,  304   b  may be omitted in some applications. 
     In normal converter operation, the gates of the top and bottom protective power FET switches would be controlled, along with the intermediate power switches, by associated clock signals CLKx as in a conventional power converter. The controlling clock signals CLKx for each set of power switches during normal conversion operation are generally non-overlapping and complementary. In the example shown in  FIG.  3   , the clock signals CLK 1 , CLK 6  for the top and bottom protective power FET switches S w   1 , S w   6  may be coupled through the respective switch controllers  302   a ,  302   b  so that the normal clocking sequence may be interrupted as needed when the FET switches S w   1 , S w   6  are to be forced to an open state. In some embodiments, the switch controllers  302   a ,  302   b  may be as simple as a single AND-gate coupled to the gates of their associated top and bottom FET power switches (through suitable voltage level shifters and drivers if needed), with a corresponding clock signal coupled to a first AND-gate input and the output of a sensing circuit coupled to a second AND-gate input. In the event of a transient or fault condition, the sensing circuit would actively set the associated FET power switch to an OFF state or to a high R ON  state. 
     The embodiment shown in  FIG.  3    provides actively controlled protection for the intermediate FET power switches in response to the one or more sensing circuits  304   a ,  304   b . Such sensing circuitry helps limit how much power is dropped across the top and bottom FET power switches by stopping power flow during prolonged transient or fault conditions and is also helpful during startup. However, by selecting high-voltage and lower saturation current characteristics for the bracketing top and bottom protective switches, fast automatic (passive) protection can be provided for the intermediate FET power switches without such active control circuitry. Saturation occurs when increasing drain-to-source voltage applied to a FET at a specific level of gate-source voltage, V GS , can no longer increase the drain-to-source current - that is, current becomes substantially constant above the V DS  saturation point. Thus, lower saturation current characteristics for the bracketing top and bottom protective switches relative to the intermediate FET power switches ensures that during a large over-current event while V IN  is connected to the power converter (i.e., Sw1 is closed), the top high-voltage FET power switch Sw1 will saturate before the intermediate FET power switches saturate, and thus the top high-voltage FET power switch Sw1 can automatically limit current to safe levels for all low-voltage switches. The designed high-voltage characteristic of the top FET power switch S w   1  ensures that it can handle the maximum expected voltage for the application. 
     Similarly, the bottom FET power switch (Sw6 in this example) can be selected so that it will saturate before the intermediate power switches saturate. This ensures that during a large over-current event while connected to a fly capacitor referenced to V IN  (e.g., when S w   6  is closed), the bottom FET power switch will saturate before the intermediate power switches saturate, and thus the bottom high-voltage FET power switch can automatically limit current to safe levels for all low-voltage switches. Notably, the power converter may continue operation since the top and bottom protective power FET switches need not necessarily be forced to an open state by the controller, as may be the case for short-duration transient or fault conditions. 
     Accordingly, top and bottom high-voltage, low-saturation power FET switches automatically protect against a transient event and enable use of low-voltage FETs for the remaining intermediate power switches. Thus, in some applications, the switch controllers  302   a ,  302   b  and sensing circuits  304   a ,  304   b  may be omitted. 
     The embodiment shown in  FIG.  3    is one example of a circuit and corresponding method for protecting a multi-level power converter that needs no more than two high-voltage FET switches while allowing all of the other power switches to be low-voltage FET switches. The protection is fast and automatic when the bracketing top and bottom high-voltage FET switches have low saturation characteristics relative to the intermediate power switches, while also being controllable by opening the protective FET switches when transients or fault conditions occur. 
     Second Embodiment 
       FIG.  4    is a block diagram of a second embodiment  400  of a multi-level converter circuit with protection circuitry in accordance with the present invention. The illustrated multi-level converter circuit is a 4-level multi-level DC-to-DC converter similar in many aspects to the multi-level converter circuit of  FIG.  3   . However, instead of requiring that the “top” and “bottom” FET switches be high-voltage FETs, a first normally-closed protective switch SP T  is added in series between V IN  and a top low-voltage FET switch (S w   1 ), and a second normally-closed protective switch SP B  is added in series between a bottom low-voltage FET switch (Sw6) and a reference voltage (e.g., circuit ground). 
     The protective switches SP T , SP B  are shown coupled to respective switch controllers  402   a ,  402   b , which may be coupled to one or more respective sensing circuits  404   a ,  404   b  that cause the switch controllers  402   a ,  402   b  to open one or both of the protective switches SP T , SP B . For example, a voltage detector coupled to V IN  may provide an “overvoltage” trigger signal to the switch controllers  402   a ,  402   b  when V IN  exceeds or falls below a set threshold voltage. In some embodiments, such a voltage detector may alone be one or both of the switch controllers  402   a ,  402   b . In some embodiments, the switch controllers  402   a ,  402   b  may be coupled to multiple sensing or measurement circuits so as to be able to open the associated protective switches SP T , SP B  in response to multiple types of transient events or fault events, such as thermal shutdown of the power converter, over-current detection through the power converter, output over-voltage and/or under-voltage detection, and/or capacitor faults (e.g., charge imbalances, open or short detections, etc.). In some embodiments, the switch controllers  402   a ,  402   b  may also respond to external supplied signals or commands that can cause the associated protective switches SP T , SP B  to open or close. 
     As noted above, the protective switches SP T , SP B  are normally-closed and thus normally always ON (conducting) - that is, the protective switches SP T , SP B  are not subject to the normal toggling of the power switches Swx. As always-ON devices, the protective switches SP T , SP B  may be easier to implement. For example, the protective switches SP T , SP B  potentially may be fabricated using any of a number of processes or technologies (e.g., as FET, bipolar, or MEMS switches), and need not be identical. The protective switches SP T , SP B  may be fabricated on the same integrated circuit (IC) die as the power switches Swx, or may be fabricated on a different IC die and co-packaged with the IC die bearing the power switches Swx, or may be off-chip discrete devices coupled to an IC die on which the power switches Swx are fabricated, such as on a printed circuit board (PCB). 
     In various embodiments, the power switches Swx in the multi-level power converter of  FIG.  4    may all be implemented as low-voltage FET switches, since triggered opening of the protective switches SP T , SP B  will generally protect the top and bottom FET power switches (S 1 , S 6  in the illustrated example) from high values of V IN  while the intermediate FET power switches (S 2 -S 5  in the illustrated example) are protected from excessive voltages not only by the protective switches SP T , SP B  but by the voltages on their associated bracketing fly capacitors C FX . 
     While illustrated switch controllers  402   a ,  402   b  are shown as distinct circuit blocks, they may be implemented as a single circuit block. The switch controllers  402   a ,  402   b  may be implemented on the same IC die as their associated protective switches SP T , SP B , and/or may be part of the overall controller for the power converter (such as part of a power converter controller like controller  108  in  FIG.  1   ). 
     When the protective switches SP T , SP B  are implemented as MOSFETs, since they are normally conducting and are not part of the switching cycle for the power switches Swx, the gate charge of the protective switches SP T , SP B  does not add to transition losses for the power converter. This can be an efficiency advantage when the power switches Swx are switching at high frequencies. 
     Note that the presence of the top protective switch SP T  provides an additional advantage during startup of the power converter. The problem of startup in-rush current can be alleviated and possibly eliminated by use of the top protective switch SP T  to provide a “soft start” mode of operation that controls in-rush current to a power converter until the fly capacitors C FX  are sufficiently charged to allow the power switches to begin normal charge transfers. For example, while the fly capacitors are first being soft-started, the top protective switch SP T  protects the power FETs from high levels of V IN  in the manner described above. In addition, a FET-based top protective switch SP T  can be actively controlled through the switch controller  402   a  to function as a fixed current source or resistor (i.e., by operating the top protective switch SP T  in saturation rather than in its linear region). Alternatively, a current source or resistor (not shown) added in parallel with the top protective switch SP T  can be used as a “bypass” current path to slowly control charging of the fly capacitors while the top protective switch SP T  continues to provide over-voltage protection. 
     Third Embodiment 
       FIG.  5    is a block diagram of a third embodiment  500  of a multi-level converter circuit with protection circuitry in accordance with the present invention. The illustrated multi-level converter circuit is a 4-level multi-level DC-to-DC converter variant of the multi-level converter circuit of  FIG.  4   . In the illustrated embodiment, the protective switches SP T , SP B  of  FIG.  4    are shown respectively implemented as FET protective switches  502 ,  504 . More specifically, protective switch SP T  is implemented as a high-voltage N-type MOSFET, while protective switch SP B  is implemented as a high-voltage P-type MOSFET. In alternative embodiments, the protective switch SP T  may be implemented as a high-voltage switch NPN bipolar junction transistor and the protective switch SP B  may be implemented as a high-voltage switch PNP bipolar junction transistor. 
     Switch controllers  506   a ,  506   b  provide a suitable control signal to the corresponding gates of the FET protective switches  502 ,  504  through respective level shifters  510 ,  514  and driver circuits  512 ,  516 . Power for the level shifters  510 ,  514  and driver circuits  512 ,  516  may be provided, for example, by a simple charge pump (not shown) coupled between V IN  and a linear regulator (not shown). In addition, the switch controllers  506   a ,  506   b  may be coupled to one or more respective sensing circuits  508   a ,  508   b  that cause the switch controllers  506   a ,  506   b  to open one or both of the FET protective switches  502 ,  504  in response to transients or fault conditions. While illustrated switch controllers  506   a ,  506   b  are shown as distinct circuit blocks, they may be implemented as a single circuit block. 
     As in the embodiment of  FIG.  4   , the top FET protective switch  502  is coupled to V IN  and is normally closed (conducting), and accordingly its nodes are not subject to any switching voltages, thereby reducing loss. Similarly, the bottom FET protective switch  504  is coupled to a reference voltage (e.g., circuit ground) and is normally closed (conducting), and accordingly its nodes are not subject to any switching voltages, again reducing loss. 
     When a fault condition occurs, the switch controllers  506   a ,  506   b  may be triggered by the sensing circuits(s)  508   a ,  508   b  and open the FET protective switches  502 ,  504 . In addition, the top FET protective switch  502  should, in general, be selected so that it will saturate before the power switches Swx saturate. This ensures that during a large over-current event while V IN  is connected to the power converter (i.e., S w   1  is closed), the FET protective switch  502  will saturate before the intermediate power FETs (S w   1 -S w   6  in this example) in the converter stack saturate, and thus the FET protective switch  502  will absorb any high voltage (up to the point of breakdown). Early entry into saturation by the FET protective switch  502  essentially limits current flow from exceeding safe levels for all low-voltage switches by keeping them operating in their linear region with very little voltage drop across them. 
     Similarly, the bottom FET protective switch  504  should be selected so that it will saturate before the power switches Swx saturate. This ensures that during a large over-current event while connected to a fly capacitor (e.g., S w   6  is closed), the FET protective switch  504  will saturate before the power FETs in the converter stack saturate, and thus the FET protective switch  504  will absorb any high voltage. 
     Accordingly, the top and bottom FET protective switches  502 ,  504 , while normally closed, may be switched to an open protective state either under active logic control when a fault condition occurs, or automatically when a transient occurs and one or both of the FET protective switches  502 ,  504  saturate. In either case, the designed high-voltage characteristic of the FET protective switches  502 ,  504  ensures that they can handle the maximum expected voltage for the application. 
     The saturation point of the FET protective switches  502 ,  504  relative to the saturation point of the low-voltage intermediate power FETs preferably should be chosen to also guarantee that the fly capacitors C FX  should be capable of maintaining their respective voltages on all their associated power switches Swx. If the low-voltage intermediate power FETs are not saturated, then the fly capacitors C FX  should be able to maintain their node voltages. Accordingly, if the fly capacitors C FX  have not fully charged (e.g., at startup), the FET protective switches  502 ,  504  absorb the difference in voltage while the fly capacitors are referenced to circuit ground (e.g., the bottom power switch S w   6  is closed, in which case FET protective switch  502  protects the intermediate power FETs) or are referenced to V IN  (e.g., the top power switch Sw1 is closed, in which case FET protective switch  504  protects the intermediate power FETs. 
     Note that when the top FET protective switch  502  is closed, then (without more) the top power switch S w   1 , when open, could be exposed to high values of V IN  or to current-induced voltage spikes. However, by adding an auxiliary protection circuit, the voltage across the open top power switch S w   1  can be clamped to a controlled level less than its breakdown voltage. In the illustrated example, the auxiliary protection circuit comprises a Zener diode Z 1  between the gate of the top FET protective switch  502  and the bottom terminal (source) of power switch S w   1  (i.e., the terminal farthest from V IN ). The Zener voltage value for Z 1  preferably approximates the breakdown voltage of power switch Sw1 (e.g., 5 V for both devices). The top FET protective switch  502  is a source follower, which means the source is a threshold voltage V TH  below the gate voltage. Thus, if the gate voltage is limited to the voltage across Z 1  when power switch S w   1  is closed, then the source of the top FET protective switch  502  is also limited to that voltage, which will clamp the voltage across power switch Sw1 to the Zener voltage Z 1  minus the V TH  of the top FET protective switch  502  (i.e., to Z1 - V TH ), and the top FET protective switch  502  will absorb any voltage drop that exceeds that level. Accordingly, the top FET protective switch  502  and the auxiliary protection circuit function as a nearly-instantaneous clamping circuit protecting the top power switch S w   1  from excessive voltages, thereby allowing the top power switch S w   1  to be implemented as a low-voltage device (e.g., a FET with a BV DS  that can be as low as the Z 1  voltage, which may be as low as V IN /3 in this example). As should be appreciated, other circuits may be used in place of the Zener diodes to perform voltage limiting. 
     A similar approach may be used to clamp the voltage across the bottom power switch (Sw6 in this example), except that the protective switch is a P-type MOSFET  504  and a Zener diode Z 2  is coupled between the gate of the bottom FET protective switch  504  and the top terminal (drain) of power switch S w   5  (i.e., the terminal farthest from circuit ground). The gate drive voltage requirements for an enhancement mode P-type MOSFET may require a negative value below circuit ground. In applications where such a voltage may not be available, a depletion mode P-type MOSFET may be employed in lieu of the P-type MOSFET  504 , since a negative drive voltage generally would not be necessary. 
     In variant embodiments, either the top FET protective switch  502  or the top power switch S w   1  can be selected to saturate first (rather than selecting the top FET protective switch  502  to saturate before the top power switch S w   1 ). If the top power switch S w   1  saturates first, the voltage across it will start to climb but when the voltage exceeds the Zener voltage of Z 1 , then the top FET protective switch  502  will begin protecting the top power switch S w   1 . Thus, Zener diode Z 1  combined with the top FET protective switch  502  allows the top power switch S w   1  to saturate without exceeding damaging voltage levels. The same concept applies to the bottom FET protective switch  504  in combination with Zener diode Z 2 , which allow bottom power switch (Sw6 in this example) to saturate without exceeding damaging voltage levels. 
     Fourth Embodiment 
       FIG.  6    is a block diagram of a fourth embodiment  600  of a multi-level converter circuit with protection circuitry in accordance with the present invention. The illustrated multi-level converter circuit is a 4-level multi-level DC-to-DC converter variant of the multi-level converter circuit of  FIG.  4   . In the illustrated embodiment, the protective switches SP T , SP B  of  FIG.  4    are shown respectively implemented by FET protective switches  602 ,  604 . More specifically, protective switch SP T  is implemented as a high-voltage P-type MOSFET, while protective switch SP B  is implemented as a high-voltage N-type MOSFET. 
     The P-type top FET protective switch  602  is shown positioned between the top power switch (S w   1 ) and the next power switch (S w   2 ) in the illustrated stack of power switches Swx. Similarly, the N-type bottom FET protective switch  604  is shown positioned between the bottom power switch (S w   6 ) and the preceding power switch (S w   5 ) in the illustrated stack of power switches Swx. A voltage source Vmax1 is coupled between V IN  and the gate of the top FET protective switch  602 . Similarly, a voltage source Vmax2 is coupled between a reference voltage (e.g., circuit ground) and the gate of the bottom FET protective switch  604  (S w   6  in this example). 
     Optional switch controllers  606   a ,  606   b  may be coupled to one or both of the voltage sources Vmax1 and Vmax2 and to one or more sensing circuits  608   a ,  608   b  that cause the switch controllers  402   a ,  402   b  to alter the voltages of Vmax1 and/or Vmax2 to open one or both of the FET protective switches  602 ,  604  in response to transients or fault conditions. In alternative embodiments, the optional switch controllers  606   a ,  606   b  may be coupled to directly control operation of the respective FET protective switches  602 ,  604 . In either case, this cascode configuration for the FET protective switches  602 ,  604  protects all of the low-voltage power switches Swx at all times, and thus the low-voltage power switches Swx may be turned OFF (opened) during a transient or a fault condition instead of turning OFF (opening) the high-voltage FET protective switches  602 ,  604 , with the same effect. This means that the FET protective switches  602 ,  604  could be permanently connected respectively to Vmax1 and Vmax2 and never toggled OFF - instead, the top power switch S w   1  and the bottom power switch (S w   6  in this example) can do the toggling. While illustrated switch controllers  606   a ,  606   b  are shown as distinct circuit blocks, they may be implemented as a single circuit block. 
     As with the embodiment of  FIG.  4   , in some embodiments, the top and bottom FET protective switches  602 ,  604  may be selected so that they will saturate before the power switches Swx saturate. This ensures that during a large over-current event while one or more of the power switches Swx is closed, the FET protective switches  602 ,  604  will saturate before the power FETs Swx in the converter stack saturate, and thus the high-voltage FET protective switches  602 ,  604  will absorb any excessively high voltage. In variant embodiments, either the top FET protective switch  602  or the top power switch S w   1  may be selected to saturate first (rather than selecting the top FET protective switch  602  to saturate before the top power switch S w   1 ). Similarly in such variant embodiments, either the bottom FET protective switch  604  or the bottom power switch (Sw6 in this example) may be selected to saturate first (rather than selecting the bottom FET protective switch  604  to saturate before the bottom power switch S w   6 ). 
     As in the embodiment of  FIG.  4   , the top FET protective switch  602  is normally closed (conducting). Similarly, the bottom FET protective switch  604  is normally closed (conducting). It might seem that the position of the FET protective switches  602 ,  604  may not provide protection for the top and bottom power switches (S w   1  and S w   6  in this example). However, the voltage sources Vmax1 and Vmax2 should be set to at least the maximum voltage that the associated power switch (S w   1  and S w   6 ) can handle (e.g., Vmax1 ≥ BV DS  for S w   1 , Vmax2 ≥ BV DS  for S w   6 ), and the top and bottom FET protective switches  602 ,  604  should be configured to handle any excess voltage above that level. For example, if the maximum expected voltage for V IN  is 15 V, and the top power switch Sw1 has a breakdown voltage of 5 V, then the voltage source Vmax1 should be set at 5 V (or at 5 V plus the threshold voltage V TH  of the switch Sw1) and the top FET protective switch  602  should be designed to withstand at least 10 V. If the actual value of V IN  exceeds Vmax1, then the top FET protective switch  602  will saturate and thus automatically limit current, thereby limiting the voltage to which power switch Sw1 is subjected. 
     The saturation point of the FET protective switches  602 ,  604  preferably should be chosen to also guarantee that the fly capacitors C FX  will be capable of maintaining their respective voltages on all their associated power switches Swx. If the low-voltage intermediate power FETs are not saturated, then the fly capacitors C FX  will be able to maintain their node voltages. Accordingly, if the fly capacitors C FX  have not fully charged (e.g., at startup), the FET protective switches  602 ,  604  absorb the difference in voltage while the fly capacitors are referenced to circuit ground (e.g., the bottom power switch Sw6 is closed). 
     Example Control Circuitry for an M-level Converter Cell 
       FIG.  7    is a block diagram of one embodiment of control circuitry  700  for an M-level converter cell  702  coupled to an output block  704  comprising an inductor L and an output capacitor C OUT  (conceptually, the inductor L also may be considered as being included within the M-level converter cell  401 ). This example control circuitry  700  is adapted from the teachings set forth in U.S. Pat. Application Serial No. 63276923, filed Nov. 8, 2021, entitled “Controlling Charge-Balance and Transients in a Multi-Level Power Converter” [Attorney Docket. No. PER-370-PROV], assigned to the assignee of the present invention, the contents of which are incorporated by reference. However, the present invention may be used in combination with other types of control circuitry for an M-level converter cell  702 . 
     The control circuitry  700  functions as a control loop coupled to the output of the M-level converter cell  702  and to switch control inputs of the M-level converter cell  702 . In general, the control circuitry  700  is configured to monitor the output (e.g., voltage and/or current) of the M-level converter cell  702  and dynamically generate a set of switch control inputs to the M-level converter cell  702  that attempt to stabilize the output voltage and/or current at specified values, taking into account variations of V IN  and output load. In alternative embodiments, the control circuitry  700  may be configured to monitor the input of the M-level converter cell  702  (e.g., voltage and/or current) and/or an internal node of the M-level converter cell  702  (e.g., the voltage across one or more fly capacitors or the current through one or more power switches). Accordingly, most generally, the control circuitry  700  may be configured to monitor the voltage and/or current of a node (e.g., input terminal, internal node, or output terminal) of the M-level converter cell  702 . The control circuitry  700  may be incorporated into, or separate from, the overall controller  104  for a power converter  100  embodying the M-level converter cell  702 . 
     A first block comprises a feedback controller  706 , which may be a traditional controller such as a fixed frequency voltage mode or current mode controller, a constant-on-time controller, a hysteretic controller, or any other variant. The feedback controller  706  is shown as being coupled to V OUT  from the M-level converter cell  702 . In alternative embodiments, the feedback controller  706  may be configured to monitor the input of the M-level converter cell  702  and/or an internal node of the M-level converter cell  702 . The feedback controller  706  produces a signal directly or indirectly indicative of the voltage at V OUT  that determines in general terms what needs to be done in the M-level converter cell  702  to maintain desired values for V OUT : charge, discharge, or tristate (i.e., open, with no current flow). 
     In the illustrated example, the feedback controller  706  includes a feedback circuit  708 , a compensation circuit  710 , and a PWM generator  712 . The feedback circuit  708  may include, for example, a feedback-loop voltage detector which compares V OUT  (or an attenuated version of V OUT ) to a reference voltage which represents a desired V OUT  target voltage (which may be dynamic) and outputs a control signal to indicate whether V OUT  is above or below the target voltage. The feedback-loop voltage detector may be implemented with a comparison device, such as an operational amplifier (op-amp) or transconductance amplifier (gm amplifier). 
     The compensation circuit  710  is configured to stabilize the closed-loop response of the feedback controller  706  by avoiding the unintentional creation of positive feedback, which may cause oscillation, and by controlling overshoot and ringing in the step response of the feedback controller  706 . The compensation circuit  710  may be implemented in known manner, and may include LC and/or RC circuits. 
     The PWM generator  712  generates the actual PWM control signal which ultimately sets the duty cycle of the switches of the M-level converter cell  702 . In some embodiments, the PWM generator  712  may pass on additional optional control signals CTRL indicating, for example, the magnitude of the difference between V OUT  and the reference voltage (thus indicating that some levels of the M-level converter cell  702  should be bypassed to get to higher or lower levels), and the direction of that difference (e.g., V OUT  being greater than or less than the reference voltage). In other embodiments, the optional control signals CTRL can be derived from the output of the compensation circuit  710 , or from the output of the feedback circuit  708 , or from a separate comparator (not shown) coupled to, for example, V OUT . One purpose of the optional control signals CTRL is for advanced control algorithms, when it may be beneficial to know how far away V OUT  is from a target output voltage, thus allowing faster charging of the inductor L if the V OUT  is severely under regulated. 
     A second block comprises an M-level controller  714 , the primary function of which is to select the switch states that generate a desired V OUT  while maintaining a charge-balance state on the fly capacitors within the M-level converter cell  702  every time an output voltage level is selected, regardless of what switch state or states were used in the past. 
     The M-level controller  714  includes a Voltage Level Selector  716  which receives the PWM control signal and the additional control signals CTRL if available. In addition, the Voltage Level Selector  716  may be coupled to V OUT  and/or V IN , and, in some embodiments, to HIGH/LOW status signals, C Fx   _H/L , from voltage detectors coupled to corresponding fly capacitors C FX  within the M-level converter cell  702 . A function of the Voltage Level Selector  716  is to translate the received signals to a target output voltage level (e.g., on a cycle-by-cycle basis). The Voltage Level Selector  716  typically will consider at least V OUT  and V IN  to determine which target level should charge or discharge the output of the M-level converter cell  702  with a desired rate. 
     The output of the Voltage Level Selector  716  is coupled to an M-level Switch State Selector  718 , which generally would be coupled to the status signals, C FX   _H/L , from the capacitor voltage detectors for the fly capacitors C FX . Taking into account the target level generated by the Voltage Level Selector  716 , the M-level Switch State Selector  718  determines which switch state for the desired output level should be best for capacitor charge-balance. The M-level Switch State Selector  718  may be implemented, for example, as a look-up table (LUT) or as comparison circuitry and combinatorial logic or more generalized processor circuitry. The output of the M-level Switch State Selector  718  is coupled to the switches of the M-level converter cell  702  (through appropriate level-shifter circuits and drivers circuits, as may be needed for a particular converter cell) and includes the switch state settings determined by the M-level Switch State Selector  718  (which selects the configuration of switches within the M-level converter cell  702  corresponding to a selected target level). 
     In general (but not always), the Voltage Level Selector  716  and the M-level Switch State Selector  718  only change their states when the PWM signal changes. For example, when the PWM signal goes high, the Voltage Level Selector  716  selects which level results in charging of the inductor L and the M-level Switch State Selector  718  sets which version to use of that level. Then when the PWM signal goes low, the Voltage Level Selector  716  selects which level should discharge the inductor L and the M-level Switch State Selector  718  sets which version of that level to use. Thus, the Voltage Level Selector  716  and the M-level Switch State Selector  718  generally only change states when the PWM signal changes (the PWM signal is in effect their clock signal). However, there may be situations or events where it is desirable for the CTRL signals to change the state of the Voltage Level Selector  716 . Further, there may be situations or events where it is desirable for the C Fx   _H/L  status signal(s) from voltage detectors coupled to the fly capacitors C Fx  within the M-level converter cell  702  to cause the M-level Switch State Selector  718  to select a particular configuration of power switch settings, such as when a severe mid-cycle imbalance occurs. In some embodiments, it may be useful to include a timing function that forces the M-level Switch State Selector  718  to re-evaluate the optimal version of the state periodically, for example, in order to avoid being “stuck” at one level for a very long time, potentially causing charge imbalances. 
     In embodiments that utilize the teachings set forth in the patent application entitled “Controlling Charge-Balance and Transients in a Multi-Level Power Converter” referenced above, the M-level controller  714  implements a control method for the M-level converter cell  702  that selects an essentially optimal switch state which moves the fly capacitors C Fx  towards a charge-balance state every time a voltage level at the L x  node is selected, regardless of what switch state or states were used in the past. Accordingly, such multi-level converter circuits are free to select a different switch state or L x  voltage level every switching cycle without a need to keep track of any prior switch state or sequence of switch states. 
     One notable benefit of the control circuitry shown in  FIG.  7    is that it enables generation of voltages in boundary zones between voltage levels, which represent unattainable output voltages for conventional multi-level DC-to-DC converter circuits. 
     In alternative unregulated charge-pumps embodiments, the feedback controller  706  and the Voltage Level Selector  716  may be omitted, and instead a clock signal CLK may be applied to the M-level Switch State Selector  718 . The M-level Switch State Selector  718  would generate a pattern of switch state settings that periodically charge balances the fly capacitors C Fx  regardless of what switch state or states were used in the past (as opposed to cycling through a pre-defined sequency of states). This ensures that if V IN  changes or anomalous evens occur, the system generally always seeks charge balance for the fly capacitors C Fx . 
     In some embodiments, the M-level Switch State Selector  718  may take into account the current I L  flowing through the inductor L by way of an optional current-measurement input  720 , which may be implemented in conventional fashion. 
     While  FIG.  7    shows a particular embodiment of control circuitry for an M-level converter cell as modified in accordance with the present invention, it should be appreciated that other control circuits may be adapted or devised to provide suitable switching signals for the switches within a converter cell. 
     Methods 
     Another aspect of the invention includes various methods for protecting a power converter. For example,  FIG.  8    is a process flow chart  800  showing a first method for protecting a power converter. The method includes: coupling a set of one or more intermediate field-effect transistor (FET) switches in series, the switches in the set having a saturation characteristic (Block  802 ); and coupling a switch to the set of one or more intermediate FET switches, the switch having a breakdown characteristic and a first saturation characteristic, the first saturation characteristic being less than the saturation characteristic of the set of one or more intermediate FET switches (Block  804 ). 
     As another example,  FIG.  9    is a process flow chart  900  showing a second method for protecting a power converter. The method includes: coupling a set of one or more intermediate field-effect transistor (FET) switches in series (Block  902 ); coupling a switch to the set of one or more intermediate FET switches, the switch having a control input for setting the switch to an open state or to a closed state (Block  904 ); and setting the switch to the closed state during normal operation of the power converter and setting the switch to the open state in response to one or more events (Block  906 ). 
     As still another example,  FIG.  10    is a process flow chart  1000  showing a third method for protecting a power converter. The method includes: coupling a set of one or more intermediate field-effect transistor (FET) switches in series (Block  1002 ); coupling a FET switch to the set of one or more intermediate FET switches, the FET switch having a control input for setting the FET switch to an open state or to a closed state (Block  1004 ); and setting the FET switch to the closed state during normal operation of the power converter and to the open state in response to one or more events (Block  1006 ). 
     As yet another example,  FIG.  11    is a process flow chart  1100  showing a fourth method for protecting a power converter. The method includes: coupling a set of one or more intermediate field-effect transistor (FET) switches in series, the set including a first FET power switch configured to be coupled to an input voltage source (Block  1102 ); coupling a FET switch between the first FET power switch and an adjacent FET power switch in the set of one or more FET power switches, the FET switch having a control input for setting the FET switch to an open state or to a closed state (Block  1104 ); and setting the FET switch to the closed state during normal operation of the power converter and to the open state in response to one or more events (Block  1106 ). 
     Additional aspects of the above method may include one or more of the following: wherein at least one switch in the set of switches has a breakdown characteristic that is less than the breakdown characteristic of the switch; and/or wherein the events are transient events or fault events. 
     Additional Control and Operational Considerations 
     It may be desirable to provide additional control and operational circuitry (or one or more shutdown procedures) that enables reliable and efficient operation of a power converter utilizing a multi-level converter cell designed in accordance with the present disclosure. For example, in a step-down power converter, the output voltage of a converter cell is less than the input voltage of the converter cell. Shutting down or disabling (e.g., because of a fault event, such as a short) a converter cell having a designed-in inductance connected to the output while the output load current is non-zero generally requires some means for discharging the inductor current. In some embodiments, a bypass switch may be connected in parallel with a designed-in inductance connected to the output of a converter cell and controlled to be open during normal operation and closed when shutting down the converter cell or if a fault event occurs. Ideally, in order to prevent transient ringing and to provide safe discharge of the inductor current, the bypass switch can be closed before disabling converter cell switching. In alternative embodiments using MOSFETs for the main switches of the converter, the inherent body diode connected between the body and drain terminals of each MOSFET can also discharge the inductor current. Details of these solutions, as well as alternative shutdown solutions, are taught in U.S. Pat. No. 10,686,367, issued Jun. 16, 2020, entitled “Apparatus and Method for Efficient Shutdown of Adiabatic Charge Pumps”, assigned to the assignee of the present invention, the contents of which are incorporated by reference. 
     Another consideration when combining converter cells in parallel is controlling multiple parallel power converters in order to avoid in-rush current (e.g., during a soft-start period for the power converters) and/or switch over-stress if all of the power converters are not fully operational, such as during startup or when a fault condition occurs. Conditional control may be accomplished by using node status detectors coupled to selected nodes within parallel-connected power converters to monitor voltage and/or current. Such node status detectors may be configured in some embodiments to work in parallel with an output status detector measuring the output voltage of an associated power converter during startup. The node status detectors ensure that voltages across important components (e.g., fly capacitors and/or switches) within the converter cell(s) of the power converters are within desired ranges before enabling full power steady-state operation of the parallel power converters, and otherwise prevent full power steady-state operation. The node status detectors may be coupled to a master controller that controls one or more of the parallel power converters using one or more common control signals. In furtherance of a master controller configuration, the parallel power converters may each report a power good signal (Pgood) when ready to leave a startup phase for full power steady-state operation. The master controller may essentially “AND” all such Pgood signals together, possibly along with one or more status signals from other circuits, such that the master controller does not enable full power steady-state operation of any the parallel power converter unless all of the parallel power converters are ready for that state. In essence, the Pgood signals from each parallel power converter are all tied together such that the parallel power converters may not transition out of startup phase until all the Pgood signals indicate that they are ready to transition to steady operation. Furthermore, if the Pgood signal changes due to a fault condition in one or more of the parallel power converters, the parallel power converters can transition from a steady state operation to an auto-restart or shutdown operation. Details of these solutions, as well as alternative shutdown solutions, are taught in U.S. Pat. No. 10,992,226, issued Apr. 27, 2021, entitled “Startup Detection for Parallel Power Converters”, assigned to the assignee of the present invention, the contents of which are incorporated by reference. 
     Another consideration in operating multi-level converter cells is attaining (i.e., pre-charging) and maintaining fly capacitor voltages that are essentially fully proportionally balanced so that all switches are subjected to a similar voltage stress, since unbalanced fly capacitors can lead to breakdown of a switch (particularly FET switches) due to exposure to high voltages. One solution to both pre-charging capacitor voltages and operational balancing of capacitor voltages in a multi-level DC-to-DC converter circuit is to provide a parallel “shadow” circuit that conditionally couples a fly capacitor to a voltage source or other circuit to pre-charge that capacitor, or conditionally couples two or more fly capacitors together to transfer charge from a higher voltage capacitor to a lower voltage capacitor, or conditionally couples a fly capacitor to a voltage sink to discharge that capacitor, all under the control of real-time capacitor voltage measurements. Each parallel “shadow” circuit may comprise a switch and a resistor coupled in parallel with a main switch that is part of a multi-level converter cell (in some cases, one switch-resistor pair may span two series-connected switches). This particular solution for pre-charging and/or balancing charge on fly capacitors is very fast, provides slow pre-charging of the fly capacitors during a pre-charge period, protects switches from in-rush current, and provides stable voltages for converter cell switches. Details of this solution, as well as alternative pre-charging and charge balancing solutions, are taught in U.S. Pat. No. 10,720,843, issued Jul. 21, 2020, entitled “Multi-Level DC-DC Converter with Lossy Voltage Balancing”, assigned to the assignee of the present invention, the contents of which are incorporated by reference. 
     Another solution to balancing capacitor voltages in a multi-level DC-to-DC converter circuit is to provide a lossless voltage balancing solution where out-of-order state transitions of a multi-level DC-to-DC converter cell are allowed to take place during normal operation. The net effect of out-of-order state transitions is to increase or decrease the voltage across specific fly capacitors, thus preventing voltage overstress on the main switches of the DC-to-DC converter. In some embodiments, restrictions are placed on the overall sequence of state transitions to reduce or avoid transition state toggling, thereby allowing each capacitor an opportunity to have its voltage steered as necessary, rather than allowing one capacitor to be voltage balanced before voltage balancing another capacitor. Details of this solution, as well as alternative charge balancing solutions, are taught in U.S. Pat. No. 10,770,974, issued Sep. 8, 2020, entitled “Multi-Level DC-DC Converter with Lossless Voltage Balancing”, assigned to the assignee of the present invention, the contents of which are incorporated by reference. 
     An additional consideration for some embodiments is enabling operation of multi-level converter cells such that voltages can be generated in boundaries zones between voltage levels. “Boundary zones” represent unattainable output voltages for conventional multi-level DC-to-DC converter circuits. In order to generate output voltages within a boundary zone, some embodiments essentially alternate (toggle) among adjacent (or even nearby) zones by setting states of the converter cell switches in a boundary zone transition pattern. For example, a 3-level DC-to-DC converter circuit may operate in Zone 1 for a selected time and in adjacent Zone 2 for a selected time. Thus, Zones 1 and 2 are treated as a single “super-zone”. More generally, in some cases, it may be useful to create super-zones using non-adjacent zones or using more than two zones (adjacent and/or non-adjacent). Details of this solution are taught in U.S. Pat. No. 10,720,842, issued Jul. 21, 2020, entitled “Multi-Level DC-DC Converter with Boundary Transition Control”, assigned to the assignee of the present invention, the contents of which are incorporated by reference. 
     Yet another consideration for some embodiments is protection of the main power switches and other components within a power converter from stress conditions, particular from voltages that exceed the breakdown voltage of such switches (particularly FET switches). One means for protecting a multi-level power converter uses at least one high-voltage FET switch while allowing all or most other main power switches to be low-voltage FET switches. 
     In power converters, particularly multi-level power converters, the power switches may be implemented with FETs, especially MOSFETs. For each power FET, a driver circuit is generally required. In addition, for some power FETs, a level shifter may be required to translate ground-referenced low-voltage logic ON/OFF signals from an analog or digital controller into a signal with the same voltage swing but referenced to the source voltage of the power FET that the signal is driving in order to charge or discharge the gate of the power FET and thereby control the conducting or blocking state of the power FET. In some applications, the functions of a level shifter and a driver circuit may be incorporated into one circuit. 
     As should be clear, the multi-level power converter embodiments described in this disclosure may be synergistically combined with the teachings of one or more of the additional control and operational circuits and methods described in this section. 
     General Benefits and Advantages of Multi-Level Power Converters 
     Embodiments of the current invention improve the power density and/or power efficiency of incorporating circuits and circuit modules or blocks. As a person of ordinary skill in the art should understand, a system architecture is beneficially impacted utilizing embodiments of the current invention in critical ways, including lower power and/or longer battery life. The current invention therefore specifically encompasses system-level embodiments that are creatively enabled by inclusion in a large system design and application. 
     More particularly, multi-level power converters provide or enable numerous benefits and advantages, including: 
     adaptability to applications in which input and/or output voltages may have a wide dynamic-range (e.g., varying battery input voltage levels, varying output voltages);   efficiency improvements on the run-time of devices operating on portable electrical energy sources (batteries, generators or fuel cells using liquid or gaseous fuels, solar cells, etc.);   efficiency improvements where efficiency is important for thermal management, particularly to protect other components (e.g., displays, nearby ICs) from excessive heat;   enabling design optimizations for power efficiency, power density, and form-factor of the power converter - for example, smaller-size multi-level power converters may allow placing power converters in close proximity to loads, thus increasing efficiency, and/or to lower an overall bill of materials;   the ability to take advantage of the performance of smaller, low voltage transistors;   adaptability to applications in which power sources can vary widely, such as batteries, other power converters, generators or fuel cells using liquid or gaseous fuels, solar cells, line voltage (AC), and DC voltage sources (e.g., USB, USB-C, power-over Ethernet, etc.);   adaptability to applications in which loads may vary widely, such as ICs in general (including microprocessors and memory ICs), electrical motors and actuators, transducers, sensors, and displays (e.g., LCDs and LEDs of all types);   the ability to be implemented in a number of IC technologies (e.g., MOSFETs, GaN, GaAs, and bulk silicon) and packaging technologies (e.g., flip chips, ball-grid arrays, wafer level scale chip packages, wide-fan out packaging, and embedded packaging).   

     The advantages and benefits of multi-level power converters enable usage in a wide array of applications. For example, applications of multi-level power converters include portable and mobile computing and/or communication products and components (e.g., notebook computers, ultra-book computers, tablet devices, and cell phones), displays (e.g., LCDs, LEDs), radio-based devices and systems (e.g., cellular systems, WiFi, Bluetooth, Zigbee, Z-Wave, and GPS-based devices), wired network devices and systems, data centers (e.g., for battery-backup systems and/or power conversion for processing systems and/or electronic/optical networking systems), internet-of-things (IOT) devices (e.g., smart switches and lights, safety sensors, and security cameras), household appliances and electronics (e.g., set-top boxes, battery-operated vacuum cleaners, appliances with built-in radio transceivers such as washers, dryers, and refrigerators), AC/DC power converters, electric vehicles of all types (e.g., for drive trains, control systems, and/or info-tainment systems), and other devices and systems that utilize portable electricity generating sources and/or require power conversion. 
     Radio system usage includes wireless RF systems (including base stations, relay stations, and hand-held transceivers) that use various technologies and protocols, including various types of orthogonal frequency-division multiplexing (“OFDM”), quadrature amplitude modulation (“QAM”), Code-Division Multiple Access (“CDMA”), Time-Division Multiple Access (“TDMA”), Wide Band Code Division Multiple Access (“W-CDMA”), Global System for Mobile Communications (“GSM”), Long Term Evolution (“LTE”), 5G, and WiFi (e.g., 802.11a, b, g, ac, ax), as well as other radio communication standards and protocols. 
     Fabrication Technologies &amp; Options 
     While embodiments of the invention have been described in the context of a switched-capacitor network, the invention is also applicable to charge pumps and inductor-based regulators. 
     Note that the top FET protective switches and the bottom FET protective switches in  FIGS.  5  and  6    can be “mixed and matched”. Thus, for example, the combination of the bottom FET protective switch  504  in  FIG.  5    and the top FET protective switch  602  in  FIG.  6    may be used to provide protection for the power switches Swx in a power converter. Similarly, the combination of the top FET protective switch  502  in  FIG.  5    and the bottom FET protective switch  604  in  FIG.  6    may be used to provide protection for the power switches Swx in a power converter. 
     Note also that for some applications the bottom protective switch need not be a high-voltage switch if startup algorithms are used that prevent the top switch from fully turning ON until it is safe to do so. For example, charging circuits may be included that will charge the fly capacitors until it is safe to turn on the top FET protective switch. 
     While a 4-level multi-level converter circuit is shown in the examples described above, the present invention may be used in conjunction with any level of multi-level converter circuit (e.g., 3-level, 5-level, etc.) having a stacked-switch and parallel fly capacitor structure similar to the circuits of  FIGS.  3 ,  4 ,  5 , and  6   , and may be adapted for use with power converters that use stacked switches which all reference a high voltage (e.g., V IN ) at one point in time and a low voltage (e.g., a reference voltage such as circuit ground) at another point in time. 
     In various embodiments of multi-level power converters, it may be beneficial to use specific types of capacitors, particularly for the fly capacitors. For example, it is generally useful for such capacitors to have low equivalent series resistance (ESR), low DC bias degradation, high capacitance, and small volume. Low ESR is especially important for multi-level power converters that incorporate additional switches and fly capacitors to increase the number of voltage levels. Selection of a particular capacitor should be made after consideration of specifications for power level, efficiency, size, etc. Various types of capacitor technologies may be used, including ceramic (including multi-layer ceramic capacitors), electrolytic capacitors, film capacitors (including power film capacitors), and IC-based capacitors. Capacitor dielectrics may vary as needed for particular applications, and may include dielectrics that are paraelectric, such as silicon dioxide (SiO 2 ), hafnium dioxide (HFO 2 ), or aluminum oxide Al 2 O 3 . In addition, multi-level power converter designs may beneficially utilize intrinsic parasitic capacitances (e.g., intrinsic to the power FETs) in conjunction with or in lieu of designed capacitors to reduce circuit size and/or increase circuit performance. Selection of capacitors for multi-level power converters may also take into account such factors as capacitor component variations, reduced effective capacitance with DC bias, and ceramic capacitor temperature coefficients (minimum and maximum temperature operating limits, and capacitance variation with temperature). 
     Similarly, in various embodiments of multi-level power converters, it may be beneficial to use specific types of inductors. For example, it is generally useful for the inductors to have low DC equivalent resistance, high inductance, and small volume. 
     The controller(s) used to control startup and operation of a multi-level power converter may be implemented as a microprocessor, a microcontroller, a digital signal processor (DSP), register-transfer level (RTL) circuitry, and/or combinatorial logic. 
     While the embodiments described above need no more than two high-voltage FETs, in alternative embodiments, one or more of the power switches Swx described above as low-voltage FETs may instead be implemented as high-voltage FETs as may be needed in particular applications. 
     The term “MOSFET”, as used in this disclosure, includes any field effect transistor (FET) having an insulated gate whose voltage determines the conductivity of the transistor, and encompasses insulated gates having a metal or metal-like, insulator, and/or semiconductor structure. The terms “metal” or “metal-like” include at least one electrically conductive material (such as aluminum, copper, or other metal, or highly doped polysilicon, graphene, or other electrical conductor), “insulator” includes at least one insulating material (such as silicon oxide or other dielectric material), and “semiconductor” includes at least one semiconductor material. 
     As used in this disclosure, the term “radio frequency” (RF) refers to a rate of oscillation in the range of about 3 kHz to about 300 GHz. This term also includes the frequencies used in wireless communication systems. An RF frequency may be the frequency of an electromagnetic wave or of an alternating voltage or current in a circuit. 
     With respect to the figures referenced in this disclosure, the dimensions for the various elements are not to scale; some dimensions have been greatly exaggerated vertically and/or horizontally for clarity or emphasis. References to orientations and directions (e.g., “top”, “bottom”, “above”, “below”, “lateral”, “vertical”, “horizontal”, etc.) are relative to the example drawings, and not necessarily absolute orientations or directions. 
     Various embodiments of the invention can be implemented to meet a wide variety of specifications. Unless otherwise noted above, selection of suitable component values is a matter of design choice. Various embodiments of the invention may be implemented in any suitable integrated circuit (IC) technology (including but not limited to MOSFET structures), or in hybrid or discrete circuit forms. Integrated circuit embodiments may be fabricated using any suitable substrates and processes, including but not limited to standard bulk silicon, high-resistivity bulk CMOS, silicon-on-insulator (SOI), and silicon-on-sapphire (SOS). Unless otherwise noted above, embodiments of the invention may be implemented in other transistor technologies such as bipolar, BiCMOS, LDMOS, BCD, GaAs HBT, GaN HEMT, GaAs pHEMT, and MESFET technologies. Monolithic IC implementation is particularly useful since parasitic capacitances and inductances generally can be kept low (or at a minimum, kept uniform across all units, permitting them to be compensated) by careful design. 
     Voltage levels may be adjusted, and/or voltage and/or logic signal polarities reversed, depending on a particular specification and/or implementing technology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletion mode transistor devices). Component voltage, current, and power handling capabilities may be adapted as needed, for example, by adjusting device sizes, serially “stacking” components (particularly FETs) to withstand greater voltages, and/or using multiple components in parallel to handle greater currents. Additional circuit components may be added to enhance the capabilities of the disclosed circuits and/or to provide additional functionality without significantly altering the functionality of the disclosed circuits. 
     Circuits and devices in accordance with the present invention may be used alone or in combination with other components, circuits, and devices. Embodiments of the present invention may be fabricated as integrated circuits (ICs), which may be encased in IC packages and/or in modules for ease of handling, manufacture, and/or improved performance. In particular, IC embodiments of this invention are often used in modules in which one or more of such ICs are combined with other circuit blocks (e.g., filters, amplifiers, passive components, and possibly additional ICs) into one package. The ICs and/or modules are then typically combined with other components, often on a printed circuit board, to form part of an end product such as a cellular telephone, laptop computer, or electronic tablet, or to form a higher-level module which may be used in a wide variety of products, such as vehicles, test equipment, medical devices, etc. Through various configurations of modules and assemblies, such ICs typically enable a mode of communication, often wireless communication. 
     Conclusion 
     A number of embodiments of the invention have been described. It is to be understood that various modifications may be made without departing from the spirit and scope of the invention. For example, some of the steps described above may be order independent, and thus can be performed in an order different from that described. Further, some of the steps described above may be optional. Various activities described with respect to the methods identified above can be executed in repetitive, serial, and/or parallel fashion. 
     It is to be understood that the foregoing description is intended to illustrate and not to limit the scope of the invention, which is defined by the scope of the following claims, and that other embodiments are within the scope of the claims. In particular, the scope of the invention includes any and all feasible combinations of one or more of the processes, machines, manufactures, or compositions of matter set forth in the claims below. (Note that the parenthetical labels for claim elements are for ease of referring to such elements, and do not in themselves indicate a particular required ordering or enumeration of elements; further, such labels may be reused in dependent claims as references to additional elements without being regarded as starting a conflicting labeling sequence).