Patent Publication Number: US-2021175635-A1

Title: Compact phased array millimeter wave communications systems suitable for fixed wireless access applications

Description:
FIELD 
     The present invention relates to communications systems and, more particularly, to communications systems that use phased array antennas. 
     BACKGROUND 
     Wireless radio frequency (“RF”) communications systems, such as cellular communications systems, WiFi networks, microwave backhaul systems and the like, are well known in the art. Some of these systems, such as cellular communication systems, operate in the “licensed” frequency spectrum where use of the frequency band is regulated so that only specific users in any given geographical region can operate in selected portions of the frequency band to avoid interference. Other systems such as WiFi operate in the “unlicensed” frequency spectrum which is available to all users, albeit typically with limits on transmit power to reduce interference. 
     Cellular communications systems are now widely deployed. In a typical cellular communications system, a geographic area is divided into a series of regions that are referred to as “cells,” and each cell is served by a base station. The base station may include baseband equipment, radios and antennas that are configured to provide two-way RF communications with fixed and mobile subscribers that are positioned throughout the cell. The base station antennas generate radiation beams (“antenna beams”) that are directed outwardly to serve the entire cell or a portion thereof. Typically, a base station antenna includes one or more phase-controlled arrays of radiating elements, which are commonly referred to as phased array antennas. 
     There has been a rapid increase in the demand for wireless communications, with many new applications being proposed in which wireless communications will replace communications that were previously carried over copper or fiber optic communications cables. Conventionally, most wireless communications systems operate at frequencies below 6.0 GHz, with some exceptions such as microwave backhaul systems and various military applications. As capacity requirements continue to increase, the use of higher frequencies is being considered for various applications, including frequencies in both the licensed and unlicensed spectrum. As higher frequencies are considered, the millimeter wave spectrum, which includes frequencies from approximately 25 GHz to as high as about 300 GHz, is a potential candidate, as there are large contiguous frequency bands in this frequency range that are potentially available for new applications. The use of cellular technology has also been contemplated for so-called “fixed wireless access” applications such as connecting cable television or other optical fiber, coaxial cable or hybrid coaxial cable-fiber optic broadband networks to individual subscriber premises over wireless “drop” links. There currently is interest in potentially deploying communications systems that operate in the 28 GHz to 60 GHz (or even higher) frequency range for such fixed wireless access applications using fifth generation (“5G”) cellular communications technology. 
     For many 5G cellular communications systems, full two dimensional beam-steering is being considered. These 5G cellular communications systems may be time division multiplexed systems where different users or sets of users may be served during different time slots. For example, each 10 millisecond period (or some other small period of time) may represent a “frame” that is further divided into dozens or hundreds of individual time slots. Each user may be assigned one or more of the time slots and the base station may be configured to communicate with different users during their individual time slots of each frame. With full two dimensional beam-steering, the base station antenna may generate small, highly-focused antenna beams on a time slot-by-time slot basis as opposed to a constant antenna beam that covers a full sector. These highly-focused antenna beams are often referred to as “pencil beams,” and the base station antenna adapts or “steers” the pencil beam so that it points at different users during each respective time slot. Pencil beams may have very high gains and reduced interference with neighboring cells, so they may provide significantly enhanced performance. 
     In order to generate pencil beams that are narrowed in both the azimuth and elevation planes, it is typically necessary to provide antennas having a two-dimensional array that includes multiple rows and columns of radiating elements. The antennas may be active antennas that have independent amplitude and/or phase control for each radiating element in the planar array (or for individual sub-groups of radiating elements). Such independent control of the amplitude and/or phase of the sub-components of an RF signal that are transmitted (and received) by each radiating element allows the radiating elements to act in coordinated fashion to generate directional pencil beam radiation patterns that may be pointed at individual users. While this technique can provide very high throughput, the provision of planar array antennas having large numbers of radiating elements with associated electronics that provide for independent amplitude and phase control may add a significant level of cost and complexity to the communications system. 
     SUMMARY 
     Pursuant to some embodiments of the present invention, millimeter wave communications systems are provided that include an RF printed circuit board structure that includes (1) a phased array antenna that includes a plurality of radiating elements and (2) a channel block that includes a plurality of active antenna channels that each feed a respective one of a plurality of sub-groups of the radiating elements. A first sub-set of the active antenna channels are completely positioned on a first side of the phased array antenna and a second sub-set of the active antenna channels each include a first portion that is on a second side of the phased array antenna, the second side being adjacent the first side. 
     In some embodiments, the second sub-set of the active antenna channels may include a total of one active antenna channel. In other embodiments, the second sub-set of the active antenna channels may include no more than two active antenna channels. 
     In some embodiments, second portions of all of the active antenna channels may extend generally in the same direction. 
     In some embodiments, the radiating elements may be arranged in rows and columns, and each active antenna channel may be connected to a respective one of the columns of radiating elements. 
     In some embodiments, each active antenna channel may include a high power amplifier and a low noise amplifier, and the high power amplifier and the low noise amplifier of a first of the active antenna channels are positioned closer to the phased array antenna than are the high power amplifier and the low noise amplifier of the second of the active antenna channels. 
     Pursuant to further embodiments of the present invention, power couplers for millimeter wave communications systems are provided that include a first 1×2 power coupler having a first input, first output and a second output, a second 1×2 power coupler having a second input that is coupled to the first output by a first transmission line segment, and a third 1×2 power coupler having a third input that is coupled to the second output by a second transmission line segment. The first transmission line segment includes a meandered delay line. 
     In some embodiments, these power couplers may further include a fourth 1×2 power coupler having a fourth input and a fifth 1×2 power coupler having a fifth input. In such embodiments, the second 1×2 power coupler may have a third output that is coupled to the fourth input by a third transmission line and a fourth output that is coupled to the fifth input by a fourth transmission line, where the third transmission line is longer than the fourth transmission line. 
     In some embodiments, the meandered delay line may comprise a co-planar waveguide meandered delay line and at least another portion of the first transmission line segment may comprise a microstrip transmission line segment. 
     In some embodiments, the first second and third 1×2 power couplers may each comprise a Wilkinson power coupler. 
     Pursuant to still further embodiments of the present invention, millimeter wave communications systems are provided that include a baseplate, an RF printed circuit board structure mounted on the baseplate, and an EMI shield cover mounted on the RF printed circuit board structure opposite the baseplate. The RF printed circuit board structure includes a phased array antenna and a plurality of active antenna channels formed therein, and the EMI shield cover includes at least a first cavity that covers a first portion of a first of the active antenna channels and a separate second cavity that covers a second portion of the first of the active antenna channels. 
     In some embodiments, the EMI shield cover may include downwardly extending walls that contact the RF printed circuit board structure, and respective lines of conductive vias may be formed in the RF printed circuit board structure underneath at least some of the downwardly extending walls of the EMI shield cover. 
     In some embodiments, a first integrated circuit chip amplifier may be mounted on the RF printed circuit board structure within the first cavity and a second integrated circuit chip amplifier may be mounted on the RF printed circuit board structure within the second cavity. 
     In some embodiments, a window may be provided between the first cavity and the second cavity. 
     Pursuant to yet additional embodiments of the present invention, millimeter wave communications systems are provided that include an RF printed circuit board structure, a first integrated circuit chip mounted on the RF printed circuit board structure, a second integrated circuit chip mounted on the RF printed circuit board structure, and an RF transmission line extending between the first and second integrated circuit chips. The RF transmission line includes a first co-planar waveguide transmission line segment adjacent the first integrated circuit chip and a microstrip transmission line segment that is between the first co-planar waveguide transmission line segment and the second integrated circuit chip. 
     In some embodiments, the RF transmission line may further include a second co-planar waveguide transmission line segment between the microstrip transmission line segment and the second integrated circuit chip. 
     Pursuant to still further embodiments of the present invention, substrate integrated waveguide filters are provided that include a printed circuit board comprising a dielectric substrate, a first metal layer on a top surface of the dielectric substrate that defines a top surface of the substrate integrated waveguide filter, a second metal layer on a bottom surface of the dielectric substrate that defines a bottom surface of the substrate integrated waveguide filter, a set of first conductive vias, each of the first conductive vias extending through the printed circuit board, the first conductive vias defining a first sidewall of the substrate integrated waveguide filter, a set of second conductive vias, each of the second conductive vias extending through the printed circuit board, the second conductive vias defining a second sidewall of the substrate integrated waveguide filter, and a set of third conductive vias that are between the first conductive vias and the second conductive vias, the third conductive vias dividing an interior of the substrate integrated waveguide filter into at least two cavities. A plurality of air-filled openings extend through the first metal layer, the dielectric substrate and the second metal layer, the air-filled openings extending through an interior of the substrate integrated waveguide filter. 
     In some embodiments, the substrate integrated waveguide filter may further include a co-planar waveguide to substrate integrated waveguide transition. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic diagram illustrating an architecture in which a millimeter wave communications system is used in a fixed wireless access application. 
         FIG. 2  is a schematic diagram of a radio unit of the millimeter wave communications system of  FIG. 1 . 
         FIG. 3  is a schematic top view of a digital board of the radio unit of  FIG. 2 . 
         FIG. 4A  is a schematic top view of an RF board of the radio unit of  FIG. 2 . 
         FIG. 413  is a cross-section taken along lines  4 B- 4 B of  FIG. 4A   
         FIG. 5A  is a schematic top view of a printed circuit board structure of the RF board of  FIGS. 4A-4B . 
         FIG. 5B  is a schematic side view of the printed circuit board structure of  FIG. 5A . 
         FIG. 6  is a schematic block diagram of the RF transmission paths between one of the ports and one of the phased array antennas on the printed circuit board structure of  FIGS. 5A-5B . 
         FIGS. 7A and 7B  are enlarged views of portions of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  that show transmission lines that include both microstrip and coplanar waveguide transmission line segments that reduce transmission line loss while maintaining low voltage standing wave ratio transitions to the integrated circuit chip pads. 
         FIGS. 8A through 8C  are enlarged views of portions of the first and fourth layers of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  that illustrate portions of a local oscillator distribution network thereof. 
         FIG. 9A  is an enlarged view of a portion of an example implementation of one of the bidirectional mixer/filter blocks the printed circuit board structure of  FIGS. 5A-5B . 
         FIG. 9B  is a schematic diagram illustrating a portion of a substrate integrated waveguide filter that includes air-filled holes that reduce the dielectric loss of the filter. 
         FIG. 10  is a plan view of a portion of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  that illustrates the staggered placement of the front-end sections of the active antenna channels and how the active antenna channels may wrap around the sides of the phased array antennas in order to reduce the overall width of the printed circuit board structure. 
         FIG. 11  is a plan view of a portion of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  that illustrates the use of power couplers having meandered delay lines and asymmetric splits. 
         FIG. 12  is a plan view of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  illustrating heat sink structures that are included therein to dissipate heat generated in the integrated circuit chips. 
         FIG. 13  is a plan view of a portion of an example implementation of the printed circuit board structure of  FIGS. 5A-5B  that illustrates via fence structures and DC bias coupling circuits that are implemented therein. 
         FIG. 14  is a plan view of an example implementation of the four phased array antennas included in the printed circuit board structure of  FIGS. 5A-5B . 
     
    
    
     DETAILED DESCRIPTION 
     Pursuant to embodiments of the present invention, millimeter wave communications systems are provided that have compact, high performance radio units. In some embodiments, the radio units include a plurality of phased array antennas that are configured to perform beamforming and operate as multi-input-multi-output (“MIMO”) antennas that simultaneously transmit multiple data streams to users. The millimeter wave communications systems according to embodiments of the present invention may be suitable for fixed wireless access applications and may support very high throughput communications at a reasonable cost. 
     Fixed wireless access applications refer to applications where the transmitters and receivers are at known, fixed locations. One proposed fixed wireless application is as a so-called “wireless drop” network that may be used as the last “leg” of a cable television network. Cable television networks are point-to-multipoint networks in which cable television, digital telephone, broadband Internet and/or other signals are transmitted between a headend facilities of a network operator/service provider and individual homes, apartment complexes, hotels, businesses, schools, government facilities and other subscriber premises (i.e., the physical locations of the subscribers to the network). These networks typically support two-way communications. In particular, “downstream” signals are transmitted from the headend facilities to the individual subscriber premises, and “upstream” signals are transmitted from the individual subscriber premises to the headend facilities. In a typical configuration, the downstream signals are transmitted over fiber optic cables to distribution points within individual neighborhoods where the optical signals are converted to radio frequency (“RF”) signals and distributed to individual subscriber premises over coaxial cable connections that typically run down individual streets. RF tap units are interposed along these coaxial cables, usually within an enclosure such as a pedestal. Each tap unit includes an input port and an output port that connect to respective first and second segments of the coaxial cable connection that runs down the neighborhood street, as well as one or more “tap ports.” The tap unit splits the RF signal that is received at an input port thereof, allowing some of the received signal energy to pass through the tap unit to the output port. The remainder of the received signal energy is split further and provided to the RF tap ports of the tap unit. So-called “drop” cables, such as, for example, coaxial drop cables, may run between each tap port of a tap unit and a point-of-entry device at each respective subscriber premise to connect each subscriber premise to the cable television network. 
     The drop cables that connect the cable television network to individual subscriber premises may be one of the most expensive parts of the outside plant of a cable television network, in terms of both initial installation and ongoing maintenance costs. In order to install a new drop cable, it is typically necessary for the service provider to send an installation crew to the site, equipped with cable burying equipment that can bury a drop cable as it is deployed and route the drop cable underneath driveways, sidewalks, fences and other pre-existing structures that are between the pedestal that houses the tap unit and a point of entry device at the subscriber premise. As the drop cables are almost always installed on privately owned real estate, it may be necessary to obtain easements before installation and to deal with complaints from property owners regarding damage to their lawns and/or shrubbery after installation is completed. The coaxial cable required for each drop may be expensive, as relatively long coaxial cable segments are typically required (e.g., 100-200 feet or more), and each coaxial cable segment only serves a single subscriber premise. Moreover, the buried coaxial cable is typically not installed in a protective conduit and hence has a limited lifetime, and also is susceptible to damage by private property owners digging on their properties to plant trees, install sprinkler systems, lay sod and the like. 
     Pursuant to embodiments of the present invention, millimeter wave wireless drop systems are provided that may be used in lieu of conventional drop cables. The wireless drop systems according to embodiments of the present invention may, in some embodiments, comprise one or more radio units that are mounted on a pole, pedestal, tower or other raised structure. Each radio unit may include one or more phased array antennas and associated electronics. In a typical implementation, three radio units may be mounted on the raised structure, with each radio unit serving a 120 degree sector in the azimuth plane in a manner similar to a sectorized cellular base station. Each radio unit may have a range of, for example, about 200-300 meters and may serve as the network gateway for a relatively large number of subscriber premises (e.g., up to 40-80 subscriber premises). 
     In an example embodiment, the millimeter wave wireless drop system may operate in the 28 GHz frequency band. The phased array antennas may be configured as beamforming antennas that can form relatively compact “pencil” antenna beams that are aimed at individual subscriber premises. These narrow antenna beams may have high levels of antenna gain, which helps offset the large free space loss that is incurred at millimeter wave frequencies. The system may be implemented as a time division multiplexed system in which the radio unit communicates with different subscribers during different time slots. The system may be a time division duplexed system where the downstream and upstream communications between the radio unit and each subscriber premise are transmitted in the same frequency band during different time slots. In some embodiments, each radio unit may include multiple phased array antennas in order to allow the radio unit to use multi-input-multi-output (“MIMO”) communications techniques. In an example embodiment each radio unit may include four phased array antennas and may transmit downstream signals to the subscriber premises using 4×MIMO techniques. 
     The radio units according to embodiments of the present invention may include a digital unit and an RF unit. Each of these units may be implemented as a multilayer printed circuit board structure. Integrated circuit chips such as controllers, optical-to-electrical, electrical-to-optical, digital-to-analog and analog-to-digital converters, power amplifiers, local oscillators, switches, diodes and the like may be mounted on the printed circuit board structures. The phased array antennas may also be implemented as part of the RF printed circuit board structure. 
     Embodiments of the present invention will now be discussed in further detail with reference to the attached drawings. 
       FIG. 1  is a schematic diagram illustrating an architecture in which a millimeter wave communications system is used in a fixed wireless access application. As shown in  FIG. 1 , a core fiber optic network  20  may connect head end facilities  10  of a network provider to individual neighborhoods and other locations. In each neighborhood, one or more millimeter wave communications systems  60  may be installed that provide wireless connections between the core fiber optic network  20  and individual subscriber premises  30  such as homes, apartments, office buildings and the like. Each millimeter wave communications system  60  may comprise one or more baseband units  80 , a power supply  90  and one or more radio units  100 . In the example of  FIG. 1 , a baseband unit  80  and a power supply  90  are mounted in a cabinet  50  near the base of a raised structure  40  such as, for example, a utility pole  40 . Three radio units  100  are shown mounted on the utility pole  40  with the phased array antennas  380  of each radio unit  100  pointed outwardly to serve a 120 degree “sector” in the azimuth plane (i.e., in a plane parallel to the plane defined by the horizon). Each radio unit  100  may transmit downstream signals to the subscriber premises  30  and receive upstream communications from the subscriber premises  30  in order to allow each subscriber premise  30  access to the core fiber optic network  20 . The baseband equipment  80  and power supply  90  may be connected to the radio units  100  by one or more cables  70  such as fiber optic and power cables. Thus, the millimeter wave communications system  60  may comprise the cables  70 , the baseband unit  80 , the power supply  90  and the radio units  100 . 
       FIG. 2  is a schematic diagram of one of the radio units  100  of the millimeter wave communications system  60  of  FIG. 1 . As shown in  FIG. 2 , the radio unit  100  includes a housing  110  that has a digital unit in the form of a digital board  200  and an RF unit in the form of an RF board  300  mounted therein. The housing  110  may comprise a metal housing and may have a removable front cover  120  that allows the digital board  200  and the RF board  300  to be installed within the housing  110 . The front cover  120  may include an opening  130  that is aligned with phased array antennas  380  (discussed below) that are included on the RF board  300  so that the phased array antennas  380  may transmit signals to, and receive signals from, the subscriber premises  30  without the front cover  120  blocking or otherwise interfering with such signals. A radome (not shown) may be provided that covers the opening  130  to protect the RF board  300  and the digital board  200  from the environment (e.g., rain, snow, insects, birds, etc.). The digital board  200  and the RF board  300  may be mounted back-to-back in the housing  110 , with the RF board  300  in front so that the phased array antennas  380  are facing outwardly. A plurality of connectors and/or connectorized cables (not shown) may be used to electrically connect the digital board  200  to the RF board  300 . 
       FIG. 3  is a schematic top view of a digital board  200  of the radio unit  100  of  FIG. 2 . As discussed above with reference to  FIG. 1 , the millimeter wave communications system  60  includes a baseband unit  80  that may be mounted, for example, at the bottom of the utility pole  40 . The baseband unit  80  is connected to the core fiber optic network  20  by a back-haul fiber optic cable  22 . A front-haul fiber optic cable  70  connects the baseband unit  80  to the digital board  200 . The front-haul fiber optic cable  70  may include a plurality of optical fibers. Assuming that separate optical fibers are provided for the uplink and the downlink, then the downlink optical fibers may be connected to optical-to-electrical modules  210  and the uplink optical fibers may be connected to electrical-to-optical modules  220 . The optical-to-electrical modules  210  may convert the fiber optic data received over the front-haul fiber optic cable  70  into 100 MHz digital baseband data. The outputs of the optical-to-electrical modules  210  may be connected to a field programmable gate array (“FPGA”)  230  that depacketizes the digital data and strips formatting therefrom. An output of the field programmable gate array  230  is connected to a digital-to-analog converter  240 . The digital-to-analog converter  240  converts the digital data stream that is received from the field programmable gate array  230  into an analog signal. The digital-to-analog converter  240  receives a clock signal from a clock generator  250  that is used to sample the analog data to produce an intermediate frequency such as 2 GHz. The 2 GHz signal is passed from the digital-to-analog converter  240  to a first transmit/receive switch  262 . The first transmit/receive switch  262  is provided to allow the radio unit  100  to operate as a time division duplexed system where different time slots are used for transmitting and receiving signals. The first transmit/receive switch  262  may be set either to feed a signal from the digital-to-analog converter  240  to a connector  260  that connects to the RF board  300  or to feed signals received at the connector  260  to an analog-to-digital converter  270 . 
     Signals received by the RF board  300  are passed to the digital board  200  through the connector  260 . The received signal is passed to the analog-to-digital converter  270  that samples the 2 GHz signal to produce an image in the first Nyquist zone (using the clock signal from the clock generator  250 ). The analog-to-digital converter  270  then digitizes the baseband signal. The digital baseband signal is passed to the field programmable gate array  230  which formats and packetizes the digital data. The data is then passed to an electrical-to-optical converter  220  that converts the digital data into an optical signal that is passed to the baseband unit  80  over the front-haul cable  70 . While not shown in  FIG. 3 , the above-described components of the digital board  200  may be replicated three additional times so that four separate 2 GHz signals may be passed simultaneously between the digital board  200  and the RF board  300 . The digital board  200  may further include a state machine or other controller  280  that generates control signals that are used to control components on the RF board  300 . 
       FIG. 4A  is a schematic top view of the RF board  300  of  FIG. 2 .  FIG. 4B  is a schematic cross-section taken along line  4 B- 4 B of  FIG. 4A . As shown in  FIGS. 4A-4B , the RF board  300  includes a baseplate  310 , a multi-piece EMI shield cover  316  and a multilayer printed circuit board structure  320 . The baseplate  310  acts as a mounting structure for the printed circuit board structure  320 , providing rigidity to RF board  300  and providing protection for the printed circuit board structure  320  and the integrated circuit chips mounted thereon. The baseplate  310  includes an integrated vapor chamber  312  on a lower surface thereof. Vapor in the vapor chamber  312  conducts heat away from the printed circuit board structure  320  to vent the heat from the RF board  300 . The vapor chamber  312  may have a very high thermal conductivity so as to provide a very low resistance thermal path for venting heat away from the RF board  300 . The baseplate  310  may include a peripheral lip (not visible in the figures) that directly contacts and supports the bottom side of the printed circuit board structure  320 . The baseplate  310  may also include a plurality of pedestals  314  that provide additional support to the printed circuit board structure  320  and which serve as low resistance thermal paths for venting heat away from the printed circuit board  320 . The pedestals  314  may be located directly underneath various high power integrated circuit chips  315  to facilitate venting heat generated by these devices. The baseplate  310  may comprise a high strength material having good thermal conductivity such as, for example, aluminum. Additional integrated circuit chips  315  may be mounted on the lower surface of the printed circuit board structure  320 . 
     The EMI shield cover  316  may comprise a metal or metal-containing structure that is used to reduce leakage of RF energy from the printed circuit board structure  320  and to block RF energy from external sources from coupling to the printed circuit board structure  320 , where such RF energy may appear as interference. By reducing leakage of RF energy, the EMI shield cover  316  may also reduce coupling between different active antenna channels on the printed circuit board structure  320 . The EMI shield cover  316  may comprise, for example, a cast or machined aluminum shield cover  320 . The EMI shield cover  316  may have a flat top surface and a plurality of downwardly-extending walls  317  that define a plurality of cavities  318  in a lower surface thereof. In the depicted embodiment, a total of three EMI shield covers  316 - 1  through  316 - 3  are provided (see  FIG. 4A ). It will be appreciated, however, that more or fewer EMI shield covers  316  may be used in other embodiments. The multilayer printed circuit board structure  320  is sandwiched between the baseplate  310  and the EMI shield covers  316 , as shown in  FIG. 4B . As is further shown in  FIG. 4A , four phased array antennas  380 - 1  through  380 - 4  are formed in the top layers of the printed circuit board structure  300 . 
     Thermal gaskets  319  may be placed on top of the high power integrated circuit chips  315 , such as the power amplifier chips, so that the thermal gaskets  319  are between the integrated circuit chips  315  and the EMI shield cover  316 . The thermal gaskets  319  may be compressed as the EMI shield cover  316  is attached to the printed circuit board structure  320 . The thermal gaskets  319  may facilitate conducting heat from the top surface of the integrated circuit chips  315  to the EMI shield cover  316 . 
       FIGS. 5A-5B  are schematic views that illustrate the multilayer printed circuit board structure  320  in greater detail. In particular,  FIG. 5A  is a schematic top view of the printed circuit board structure  320  that illustrates the layout of the top metallization layer thereof in greater detail, while  FIG. 5B  is a schematic side view that illustrates the layer structure of the printed circuit board structure  320 . 
     Referring first to  FIG. 5B , the printed circuit board structure  320  comprises a multi-layer printed circuit board having ten metallization layers  324 - 1  through  324 - 10  that are separated by nine dielectric layers. As will be discussed in detail herein, surface mount components may be mounted on the top and bottom metallization layers  324 - 1  and  324 - 10 . The dielectric layers include core dielectric layers  326 - 1  through  326 - 5  and adhesive dielectric layers  328 - 1  through  328 - 4 . The core dielectric layers  326  may comprise standard printed circuit board materials such as, for example, Taconic TSM-DS3, FR4, Arlon AD3003A, or Rogers RO3003 printed circuit board substrate materials. The metallization layers  324  may be patterned metal layers that are formed on the top and bottom surfaces of the core dielectric layers  326  using, for example, conventional printed circuit board fabrication techniques. Thus, a total of five so-called “double-layer” printed circuit boards  322  (i.e., printed circuit boards that comprise a core dielectric layer  326  with metal layers  324  on each side thereof) may be used to form the printed circuit board structure  320 . The adhesive dielectric layers  328  may be used to adhere adjacent ones of the double-layer printed circuit boards  322  together to form the multilayer printed circuit board structure  320 . The adhesive dielectric layers  328  may comprise, for example, a “prepreg” material such as a fiberglass material or other composite fiber material that is pre-impregnated with a thermoset polymer matrix material (e.g., an epoxy resin) and a curing agent. The prepreg material becomes flowable when heated and then acts as an adhesive to bind the fibers together and to other components such as the printed circuit boards  322 . 
     Referring now to the schematic view of  FIG. 5A , the printed circuit board structure  320  includes four input/output ports  321 - 1  through  321 - 4 . Each input/output port  321  may be connected to a respective one of the connectors  260  on the digital board  200  by, for example, a coaxial cable. A plurality of RF paths extend between each input/output port  321  and a respective one of the phased array antennas  380 . Thus, the printed circuit board structure  320  may simultaneously transmit four different signals through the respective four phased array antennas  380 . These four signals may be transmitted to a single user or to multiple users during any given time slot in the time division multiplexing frame structure.  FIG. 5A  is rotated ninety degrees from the orientation that the printed circuit board structure  320  will have when the printed circuit board structure  320  is mounted for normal use so that the figure may be enlarged on the page. Thus, it will be appreciated that the printed circuit board structure  320  will be rotated ninety degrees from this orientation when mounted for use. 
     Each transmit/receive path that extends from an input/output port  321  to a respective one of phased array antennas  380  includes a bidirectional mixer/filter block  330 , a power coupler  350 , and a channel group  360 , each of which are implemented on the top metallization layer  324 - 1  of the printed circuit board structure  320 . 
     Each bidirectional mixer/filter block  330  may include a mixer that performs up-conversion on intermediate frequency signals that are to be transmitted and that performs down-conversion (to an intermediate frequency) on received RF signals. This mixer is also referred to herein as an upconverter and/or as a downconverter. In an example embodiment, each mixer may be a subharmonic mixer that uses a 13 GHz local oscillator signal to up-convert 2 GHz intermediate frequency signals to 28 GHz for transmission and to down-convert received 28 GHz signals to 2 GHz. Each bidirectional mixer/filter block  330  also includes a bandpass filter that removes unwanted intermodulation products that are generated by the mixer as well as other out-of-band noise components. The design and operation of example embodiments of the mixers and filters will be discussed in greater detail below with reference to  FIGS. 6 and 9A-9B . 
     Each power coupler  350  receives (for signals flowing in the transmit direction) the 28 GHz signal output by one of the bidirectional mixer/filter blocks  330  and splits this signal into eight separate sub-components. In some embodiments, the power coupler  350  may split the power of the 28 GHz signal into eight sub-components that have equal power, although embodiments of the present invention are not limited thereto. Each eight-way power coupler  350  may be implemented, for example, as a series of 1×2 power couplers that split an RF signal to be transmitted into eight sub-components that are passed to the eight columns  386  of an associated phased array antenna  380  and which combine eight sub-components of an RF signal received at the phased array antenna  380  into a composite received RF signal. The eight outputs of each power coupler  350  are coupled to a respective one of the channel groups  360 . An example implementation of one of the power couplers  350  will be described in greater detail below with reference to  FIG. 11 . 
     Each channel group  360  passes the sub-components of an RF signal received from one of the power couplers  350  to one of the phased array antennas  380 . Each channel group  360  includes eight active antenna channels  362 . Each of the eight active antenna channels  362  included in a channel group  360  is connected to a respective one of the eight columns  386  of the phased array antennas  380 . Each active antenna channel  362  receives the eight sub-components of the RF signal output by the power coupler  350 , modifies the amplitude and/or phase of these signals for purposes of power balancing and beam steering in the azimuth plane, amplifies the sub-components and passes the amplified sub-components of the RF signal to a respective one of the columns  386  of the phased array antenna  380  for transmission. Thus, each active antenna channel  362  may be used to independently adjust the amplitude and/or phase of a respective sub-component of an RF signal. Operation of the channel groups  360  will be discussed in greater detail below with reference to  FIG. 6 . 
     Each phased array antenna  380  comprises an 8×8 array of radiating elements  382 . In the depicted embodiment, each radiating element  382  may comprise a stacked patch radiating element. Each phased array antenna  380  may be implemented using any of the phased array antennas described in U.S. Provisional Patent Application Ser. No. 62/573,749, entitled Broadband Stacked Patch Radiating Elements and Related Phased Array Antennas, Attorney Docket No. 9833-1414-PR, filed Oct. 18, 2017, the content of which is incorporated herein by reference as if set forth in its entirety. As shown in  FIG. 5A , each phased array antenna  380  includes sixty-four radiating elements  382  that are arranged in eight rows  384  and eight columns  386 . Since, as noted above, the printed circuit board structure  320  is rotated ninety degrees in  FIG. 5A , the columns  386  extend horizontally in  FIG. 5A  while the rows  384  extend vertically. The provision of eight radiating elements  382  in each row  384  and column  386  of the phased array antennas  380  allows the phased array antennas  380  to generate antenna beams that may be significantly narrowed in both the azimuth and elevation planes. 
     As noted above, each phased array antenna  380  is fed by eight active antenna channels  362 . As will be discussed in greater detail below, an active antenna channel  362  is a channel that receives a sub-component of an RF signal that is to be transmitted (or a received sub-component of an RF signal) and passes the RF signal through an adjustable phase shifter and/or a component such as a variable attenuator that can adjust a magnitude of the RF signal so that each channel may independently adjust the magnitude and/or phase of the sub-component of an RF signal passed therethrough. Thus, the sub-components of an RF signal provided to each radiating element  382  in one of the rows  384  of radiating elements  382  in a first of the phased array antennas  380  may have an independently set magnitude and/or phase. This capability allows the phased array antenna  380  to steer the antenna beam in the azimuth plane. 
     While each row of each phased array antenna  380  is fed by a different active antenna channel  362 , each column  386  of the phased array antenna  380  is fed by the same active antenna channel  362 . Thus, each radiating element  382  in a given column  386  receives the same sub-component of the RF signal. Since the amplitudes and/or phases of the sub-component of the RF signal that are fed to each radiating element  382  in a column  386  are not independently adjustable, the phased array antenna  380  cannot perform beam steering in the elevation plane. Each phased array antenna  380 , however, may be designed to have a switched elevation beamwidth. Techniques for implementing such elevation beamwidth switching are disclosed in U.S. Provisional Patent Application Ser. No. 62/506,100, entitled  Phased Array Antennas Having Switched Elevation Beamwidths and Related Methods,  Attorney Docket No. 9833-1221-PR, filed May 15, 2017, and in U.S. Provisional Patent Application Ser. No. 62/522,859, having the same title, Attorney Docket No. 9833-1221-PR2, filed on Jun. 21, 2017, the entire content of each of which is incorporated herein by reference as if set forth in its entirety. 
     As described in the above-identified applications, one or more switches such as, for example, PIN diodes, may be interposed along each transmission line that connects an active antenna channel  362  to the radiating elements  382  in a column  386  of a phased array antenna  380 . When these switches are all in the OFF (high impedance) state, all of the radiating elements  382  in a column  386  are fed the sub-component of an RF signal, and the phased array antenna  380  may generate an antenna beam having a relatively narrow beamwidth in the elevation plane. Under these circumstances, the antenna beam may be pointed towards the far edge of the region (“cell”) covered by the phased array antenna  380 . In contrast, when one of the switches is in the ON (low impedance) state, the radiating elements  382  along the column  386  that are after the switch are effectively removed from the column  386 . When this occurs, the elevation beamwidth of the phased array antenna  380  is increased, allowing the phased array antenna  380  to generate an antenna beam that will cover subscribers that are closer to the radio unit  100  (i.e., not near the edge of the cell) or which are at greater elevation angles (e.g., subscribers in a multi-story building). While the gain of the antenna beam is reduced when elevation beamwidth switching is used to increase the elevation beamwidth, such elevation beamwidth switching is typically performed so that the antenna beam will cover subscribers that are closer to the radio unit  100 , and hence experience less free space loss. 
     Finally, as further shown in  FIG. 5A , the printed circuit board structure  320  includes first and second customizable programmable logic device (“CPLC”) circuits  390 , a 13 GHz local oscillator (“LO”) synthesizer circuit  392 , and DC power and control connectors  394 . 
     The first and second customizable programmable logic device circuits  390  are also implemented on the bottom metallization layer  324 - 1  of printed circuit board structure  320 . The customizable programmable logic device circuits  390  may be used to generate control signals that are passed to various integrated circuit chips  315  and other components mounted on the printed circuit board structure  320  such as, for example, variable attenuators, adjustable phase shifters, high power amplifiers, low noise amplifiers, switches and the like. Each customizable programmable logic device circuit  390  may decode control instructions received from the digital board  200  of the millimeter wave communications system  100  (via, for example, a high speed serial bus) and generate the control signals therefrom. 
     The 13 GHz local oscillator synthesizer circuit  392  is an integrated circuit chip that is mounted on the top metallization layer  324 - 1  of printed circuit board structure  320 . The 13 GHz local oscillator synthesizer circuit  392  consists of a phase locked loop controlling a voltage oscillator that generates a 13 GHz signal that is phase locked to a local timing reference and that is used as a local oscillator signal by the mixers in the bidirectional mixer filter blocks  330  for purposes of generating the 26 GHz signals that are used for up-conversion and down-conversion. A 13 GHz local oscillator reference signal is used to avoid the need to generate a 26 GHz local oscillator reference signal due to the high losses that would be associated with distributing a 26 GHz local oscillator reference signal to the four bidirectional mixer/filter blocks  330  that are located near the four corners of the printed circuit board structure  320 . The mixers may be implemented as sub-harmonic mixers that internally double a received local oscillator signal to convert the 13 GHz local oscillator signal into a 26 GHz signal. 
     DC power and control connectors  394  are mounted on the back side of the printed circuit board structure  320 . The DC power and control connectors  394  may be connected to mating connectors on the digital board  200  to supply DC power and control signals to the RF board  300 . 
       FIG. 6  is a block diagram of one of the bidirectional mixer/filter blocks  330 , one of the power splitters  350 , one of the channel groups  360 , and one of the phased array antennas  380  of the printed circuit board structure  320  of  FIGS. 5A-5B . As shown in  FIG. 6 , an intermediate frequency signal may be input to the printed circuit board structure  320  from the digital board  200  through an input/output port  321 . The intermediate frequency signal may comprise, for example, a 2 GHz analog data signal. It will be appreciated, however, that any suitable intermediate frequency may be used, or that baseband data may be supplied to the printed circuit board structure  320  in other embodiments 
     The 2 GHz signal may be received at input/output port  321  via a cabling connection (not shown) from the connector  260  on the digital board  200 . The 2 GHz signal is passed from input/output port  321  to an up/down converter  334  that multiplies the 2 GHz signal by a local oscillator signal to up-convert the 2 GHz signal. The up/down converter  334  may be fed by a local oscillator  336  that generates, for example, a 13 GHz signal. As noted above, the up/down converter  334  may double the 13 GHz local oscillator signal to generate a 26 GHz oscillation signal before multiplying the oscillation signal with the 2 GHz data signal to generate a 28 GHz transmit signal. This 28 GHz signal may be output by the up/down converter  334  to a first circulator  338  (or, alternatively, a transmit/receive switch). The first circulator  338  routes the 28 GHz signal to an amplifier  340  that increases the signal level to maintain an acceptable signal-to-noise ratio. The output of the amplifier  340  is fed to a second circulator  342  (or, alternatively, another transmit/receive switch) which feeds the signal to a filter  346 . 
     With respect to upstream signals, RF signals received by the phased array antenna  380  may be passed from the filter  346  to the second circulator  342 . The second circulator  342  passes such signals to a low noise amplifier  348 . The low noise amplifier  348  increases the level of the received signal to maintain an acceptable signal-to-noise ratio. The received signal is then passed through the first circulator  338  to the up/down converter  334 , which uses the local oscillator signal to downconvert the received signal to an intermediate frequency (e.g., 2.0 GHz). This downconverted signal is passed to the input/output port  321  where it is coupled to the digital board  200 . 
     The filter  346  may comprise a bandpass filter that filters out intermodulation products and local oscillator leakage generated at the up/down converter  334  and any other unwanted signals or noise. For example, the filter  346  may comprise a 28 GHz bandpass filter. The filtered 28 GHz signal output by filter  346  is passed to one of the 1×8 power couplers  350 . The power coupler  350  splits the RF signal that is to be transmitted into eight sub-components (which may or may not have equal amplitudes depending upon the design of the power coupler  350 ). Each sub-component is passed through an output leg  352  of power coupler  350  to a respective one of the active antenna channels  362  of the channel block  360 . 
     Each active antenna channels  362  forms a transmit path and a receive path between one of the outputs  352  of power coupler  350  and one of the columns  386  of the phased array antenna  380 . Each active antenna channels  362  includes a second transmit/receive switch  364  and a third transmit/receive switch  372  that are used to route signals along either a transmit path or a receive path. The transmit path includes a variable attenuator  366 , a variable phase shifter  368  and a high power amplifier  370  that are arranged in series between the second transmit/receive switch  364  and the third transmit/receive switch  372 . The variable attenuator  366  may be configured to reduce the magnitude of the sub-component of the RF signal supplied thereto by an amount determined by a control signal provided to the variable attenuator  366 . The variable attenuator  366  may comprise, for example, a switched attenuator circuit that has a plurality of different selectable attenuation values. The variable phase shifter  368  may be used to modify the phase of the sub-component of the RF signal. The variable phase shifter  368  may comprise, for example, an integrated circuit chip that may adjust the phase of a millimeter wave signal input thereto. A control signal supplied to the variable phase shifter  368  may select one of a plurality of phase shifts. The high power amplifier  370  may amplify the sub-component of the RF signal to an appropriate transmit level. The amplified sub-component of the RF signal is then passed to a column  386  of the phased array antenna  380  for over the air transmission. A splitter/combiner network (not shown) may further split the RF signal to pass a portion thereof to some or all of the radiating elements  382  included in the column  386 . 
     When operating in receive mode, a millimeter wave signal (e.g., a 28 GHz signal) may be received at some or all (depending upon the elevation beam switching mode) of the eight radiating elements  382  of the column  386  of the phased array antenna  380 . The above-mentioned splitter/combiner network (not shown) may combine the sub-components of the received signal and pass the combined received signal through the third transmit/receive switch  372  to a receive path of the active antenna channel  362 . The receive path includes a low noise amplifier  374 , a variable phase shifter  376  and a variable attenuator  378 . The low noise amplifier  374  amplifies the received signal and passes it to the adjustable phase shifter  376 , which may adjust a phase of the received signal. The output of the variable phase shifter  376  is passed to the variable attenuator  378  that may be used to reduce the magnitude of the received signal. The output of the variable attenuator  378  is passed to the second transmit/receive switch  364 , which passes the signal to the power coupler  350  which combines the RF signals received at each of the eight columns  386  of phased array antenna  380 . 
     While the above discussion only describes one of the active antenna channels  362 , it will be appreciated that the other active antenna channels  362  may operate in the same manner as discussed above. As will be discussed below, in some embodiments an RF integrated circuit beamforming chip may be used to implement the transmit path and receive path variable attenuators  366 ,  378  and phase shifters  368 ,  376  and the second transmit/receive switch  364 . 
     The phased array antenna  380  is implemented as an 8×8 array of radiating elements  382 . As the phased array antenna  380  has already been discussed in detail above (including in the co-pending provisional applications that have been incorporated herein by reference), further description thereof will be omitted here. 
       FIGS. 7A and 7B  are enlarged views of two portions of an example implementation of the printed circuit board structure  320  of  FIGS. 5A-5B  that illustrate transmission lines that include both microstrip and coplanar waveguide transmission line segments that reduce transmission line loss while maintaining low voltage standing wave ratio transitions to integrated circuit chip pads. Referring first to  FIG. 7A , an example implementation of portions of two of the active channel paths  362  are shown.  FIG. 7A  shows the “front end” portions of two of the active channel paths  362  (i.e., the portions of the active antenna channels  362  that are closest to the phased array antenna  380  along the RF transmission path). The front end portion of each active antenna channel  362  includes the third transmit/receive switch  372 , the high power amplifier  370  and the low noise amplifier  374 , each of which are implemented as an integrated circuit chip. Each integrated circuit chip  370 ,  372 ,  374  includes a number of RF input/output pads  400  around the periphery thereof that are used to input/output RF signals as well as input/output pads  402  for receiving power and control signals. Because of the narrow spacing between adjacent ones of these input/output pads  400 ,  402  it may not be possible to connect a conventional microstrip transmission line to the RF input/output pads  400 , at least while using reasonably priced printed circuit board materials and transmission line widths suitable for obtaining good impedance matches between the transmission lines and the integrated circuit chips  370 ,  372 ,  374 . 
     As shown in  FIG. 7A , the RF transmission lines  410  that extend between the integrated circuit chips  370 ,  372 ,  374  and that extend from the integrated circuit chips  370 ,  372 ,  374  to other regions of the printed circuit board structure  320  may be implemented using a combination of microstrip transmission line segments  420  and co-planar waveguide transmission line segments  430 . The microstrip transmission line segments  420  may be relatively low loss. However, as can be seen in  FIG. 7A , the microstrip transmission line segments  420  are also wider than the co-planar waveguide transmission line segments  430  and hence it may be difficult to directly connect a microstrip transmission line segment  420  directly to some of the RF input/output pads  400  on the integrated circuit chips  370 ,  372 ,  374 . 
     As known in the art, a coplanar waveguide transmission line refers to a printed circuit board based transmission line structure that includes a conductive track that is formed on a first side of a dielectric substrate and a ground plane that is formed on a second opposed side of the dielectric substrate. A pair of ground (return) conductors are formed on either side of the conductive track on the first side of the dielectric substrate, and hence are co-planar with the conductive track. The return conductors are separated from the conductive track by respective small gaps that typically have unvarying widths along the length of the co-planar waveguide transmission line. Metal-filled ground vias  432  are provided that connect the return conductors to the ground plane on the second side of the dielectric substrate. 
     As shown in  FIG. 7A , pursuant to embodiments of the present invention, various of the RF transmission lines  410  may be implemented using both microstrip and co-planar waveguide transmission line segments  420 ,  430 . The co-planar waveguide transmission line segments  430  may connect to the RF input/output pads  400 , while the microstrip transmission line segments  420  may be used in regions remote from the integrated circuit chips where there is additional room for wider transmission line segments. The transitions between the co-planar waveguide transmission line segments  430  and the microstrip transmission line segments  420  may be designed to have a substantially constant impedance to reduce or minimize transmission loss, and the co-planar waveguide transmission line segments  430  may have an improved impedance match with the integrated circuit chips to provide improved return loss performance. 
       FIG. 7B  illustrates the use of RF transmission lines having mixed co-planar waveguide transmission line segments  430  and microstrip transmission line segments  420  in the “rear end” portions of the active antenna channels  362  that include the variable attenuators  366 ,  378 , variable phase shifters  368 ,  376  and second transmit/receive switch  364 . In the depicted embodiments, a single beamforming RF integrated circuit chip (“RFIC”)  365  is used to implement the transmit and receive path variable attenuators  366 ,  378 , the transmit and receive path variable phase shifters  368 ,  376  and the second transmit/receive switch  364 . Two of the three RF transmission lines  410  that connect to the beamforming RFIC  365  are implemented as microstrip transmission lines  420  that have a co-planar waveguide transmission line segment  430  connected thereto that connect to the RF input/output pads  400  to provide a good impedance match thereto. The third transmission line  410  that connects to the beamforming RFIC  365  is implemented as a co-planar transmission line  430  that directly connects the beamforming RFIC  365  to one of two additional attenuators  378  that are provided along the receive path. 
       FIGS. 8A-8C  are three plan views of portions of the first and fourth layers of an example implementation of the printed circuit board structure  320  of  FIGS. 5A-5B  that illustrate portions of a local oscillator distribution network that are used to distribute a 13 GHz local oscillator signal from the local oscillator synthesizer  336  to each of the four mixers  334 . 
       FIG. 8A  is a view of a portion of the top metallization layer  324 - 1  of the example implementation of the printed circuit board  320 . As shown in  FIG. 8A , a Wilkinson power divider  450  is implemented on patterned metal layer  324 - 1 . As discussed above, a 13 GHz local oscillator synthesizer integrated circuit chip  392  (see  FIG. 5A ) is mounted on the top metallization layer  324 - 1  of printed circuit board structure  320 . The 13 GHz local oscillator signal that is generated by the local oscillator synthesizer integrated circuit chip  392  is amplified by an amplifier. Wilkinson power divider  450  is used to split the amplified local oscillator signal into two equal parts.  FIG. 8B  is a plan view of stripline transmission lines patterned metallization layer  324 - 3  of the example implementation of the printed circuit board  320 , with the metal traces on patterned metal layer  324 - 1  that form the Wilkinson power divider  450  superimposed thereon to provide perspective. As shown in  FIG. 8B , the inputs and outputs of the Wilkinson power divider  450  connect to patterned metallization layer  324 - 3  through vertical transitions. Stripline transmission line segments  470 - 1 ,  470 - 2 ,  470 - 3  are formed in the interior of the printed circuit board structure  320 . In particular, conductive tracks  472  are formed on metallization layer  324 - 4 . Conductive vias  474  are formed through the printed circuit board structure  320  along either side of the conductive tracks  472 . Ground planes (not visible in the figures) are provided on metallization layers  324 - 2  and  324 - 4 , so that the conductive tracks  472  are each enclosed by the ground planes and the conductive vias  474  to complete the stripline transmission line segments  472 . The stripline transmission line segments  472  are used because they have low radiated loss and because they prevent multiple transmission line crossovers that would otherwise be needed if routed on the top metalization layer. 
     Each stripline transmission line segments  470  is connected to an input of a respective Wilkinson power divider  480 - 1 ,  480 - 2  on metallization layer  324 - 3 .  FIG. 8C  is a plan view of a small portion of patterned metallization layer  324 - 4  of the example implementation of the printed circuit board  320 . The four (total) outputs of the Wilkinson power divider  480 - 1 ,  480 - 2  provide the 13 GHz local oscillator signal to the four upconverter/downconverters  334  that are discussed above with reference to  FIG. 6 . 
     As noted above, vertical transitions are included in the printed circuit board structure  320  that transition the input and outputs of the Wilkinson power divider  450  between the first patterned metallization layer  324 - 1  and the fourth patterned metallization layer  324 - 4 . These vertical transitions may be implemented, for example, using any of the vertical transition structures disclosed in U.S. Provisional Patent Application Ser. No. 62/573,244, titled  Vertical Transitions for Microwave and Millimeter Wave Communications Systems Having Multi - Layer Substrates,  Attorney Docket No. 9833-1421-PR, filed Oct. 17, 2017, the entire content of which is incorporated herein by reference. 
       FIG. 9A  is an enlarged view of a portion an example implementation of the printed circuit board structure  320  of  FIGS. 5A-5B  illustrating one possible implementation of the bidirectional mixer/filter blocks  330 . 
     As shown in  FIG. 9A , the bidirectional mixer/filter block  330  includes an up/down converter  334  that multiplies the 2 GHz data signal by a local oscillator signal to up-convert the 2 GHz data signal to 28 GHz. The up/down converter  334  receives a 13 GHz local oscillator signal from one of the Wilkinson power dividers  480  discussed above. The up/down converter  334  generates a 26 GHz signal from the 13 GHz local oscillator signal and multiplies the 2 GHz signal by the 26 GHz signal to up-convert the 2 GHz signal to 28 GHz signal for transmission. 
     The 28 GHz signal output from the up/down converter  334  is passed to a first circulator  338 . The first circulator  338  routes the 28 GHz signal to an amplifier  340  that increases the signal level to maintain an acceptable signal-to-noise ratio. The output of the amplifier  340  is fed to a second circulator  342  which feeds the signal to a filter  346 . Signals received by the phased array antenna  380  are passed in the reverse direction from the second circulator  342  to the up/down converter  334 , except that the second circulator  342  routes the received signals through amplifier  348 , which increases the level of the received signal to maintain an acceptable signal-to-noise ratio. 
     As is further shown in  FIG. 9A , the second circulator  342  connects to the filter  346  through a microstrip transmission line  600 . The end of the microstrip transmission line  600  that terminates into the filter  346  transitions first into a co-planar waveguide segment  610  and then to a co-planar waveguide to substrate integrated waveguide transition  620 . The co-planar waveguide to substrate integrated waveguide transition  620  performs impedance matching to provide a low loss transition between the two different types of transmission line structures. The filter  346  is implemented as a four cavity substrate integrated waveguide filter. As known to those of skill in the art, a substrate integrated waveguide transmission line is a waveguide structure that is formed in a multi-layer substrate (e.g., a printed circuit board) that includes a dielectric substrate with upper and lower metal layers formed thereon and two rows of conductive posts (e.g., metal-plated or metal-filled posts) that connect the upper metal layer to the lower metal layer. The combination of the two metal layers and the two rows of metal posts may define a rectangular waveguide structure in the dielectric substrate that RF signals may be transmitted through. 
     The filter  346  may filter out intermodulation products generated at the up/down converter  334  and any other unwanted signals or noise. For example, the filter  346  may comprise a bandpass filter. The filter  346  may be tuned to pass the desired (28 GHz) signals while filtering out signals and noise in other frequency bands by varying the width and/or length of each cavity  630  thereof. As described above, the mixer  334  multiplies a 2 GHz intermediate frequency signal with a 26 GHz local oscillator signal that is generated by doubling a 13 GHz local oscillator signal in the mixer  334 . As a result, intermodulation products may be created at 11 GHz, 15 GHz and 24 GHz along with the desired 28 GHz signal, and 13 GHz and 26 GHz signals may also couple onto the transmission path. The substrate integrated waveguide filter  346  may filter out these intermodulation products along with other out-of-band noise. 
     In some embodiments, holes may be drilled through the printed circuit board structure  320  that extend through the filter  346 . These holes may be filled with air. If properly designed in terms of size and spacing, the holes will not result in material leakage of RF energy flowing through the filter  346 . The holes, however, may reduce the overall dielectric constant within the interior of the substrate integrated waveguide filter  346 , which may advantageously reduce transmission losses.  FIG. 9B  is a schematic diagram illustrating an example implementation of a substrate integrated waveguide filter  800  that includes such air-filled holes that reduce the dielectric loss of the filter. 
     As shown in  FIG. 9B , the substrate integrated waveguide filter  800  is formed in a printed circuit board  802  that has a dielectric substrate  810 , a first metal layer  820  that is formed on a top surface of the dielectric substrate  810  and a second metal layer  830  that is formed on a bottom surface of the dielectric substrate  810 . The first metal layer  820  may define a top surface of the substrate integrated waveguide filter  800  and the second metal layer  830  may define a bottom surface of the substrate integrated waveguide filter  800 . 
     The substrate integrated waveguide filter  800  further includes a set of first conductive vias  840 , each of which extends through the printed circuit board  802 . The first conductive vias  840  may extend in a row to define a first sidewall  860  of the substrate integrated waveguide filter  800 . The substrate integrated waveguide filter  800  also includes a set of second conductive vias  842 , each of which extends through the printed circuit board  802 . The second conductive vias  842  may extend in a row to define a second sidewall  870  of the substrate integrated waveguide filter  800 . A set of third conductive vias  844  are provided that are between the first conductive vias  840  and the second conductive vias  842 . The third conductive vias  844  may divide an interior of the substrate integrated waveguide filter  800  into at least two cavities  880 ,  882 . Additionally, a plurality of air-filled openings  850  extend through the first metal layer  820 , the dielectric substrate  810  and the second metal layer  830 , the air-filled openings  850  extending through an interior of the substrate integrated waveguide filter  800 . While not shown in  FIG. 9B , the substrate integrated waveguide filter  800  may further include a co-planar waveguide to substrate integrated waveguide transition  620 . 
     Pursuant to additional embodiments of the present invention, techniques are provided for reducing the overall surface area on the printed circuit board structure required to implement the active antenna channels  362 . In particular,  FIG. 10  is a schematic block diagram of the printed circuit board structure  320  of  FIGS. 5A-5B  that illustrates how the front-end sections  363  of the active antenna channels  362  may be staggered with respect to each other and how some of the active antenna channels  362  may wrap around the sides of the phased array antennas  380  to further reduce the width of the printed circuit board structure  320 . 
     As shown in  FIG. 10 , the placement of the front end sections  363  of the active antenna channels  362  are staggered, with some of the front end sections  363  being placed in close proximity to the phased array antennas  380 , while others are located at significantly larger distances from the phased array antennas  380 . This arrangement allows the active antenna channels  362  to be placed closer together, reducing the width of the printed circuit board structure  320  and distributing the power amplifier heat load over a larger surface area. As can also be seen in  FIG. 10 , the front end sections  363  of the two active antenna channels  362  at the end of each channel group  360  wrap around the sides of the phase array antennas  380 . This allows for further reduction in the width of the printed circuit board structure  320 . 
     As is further shown in  FIG. 10 , walls  650  of conductive vias  652  are formed through the printed circuit board structure  320 . The walls  650  of conductive vias  652  may be formed in the locations where the EMI shield cover  316  contacts the printed circuit structure  320  so that the walls  650  of conductive vias  652  extend underneath the downwardly-extending walls  317  of the EMI shield cover  316 . The walls  650  of conductive vias  652  provide additional isolation between adjacent active antenna channels  362 . 
     Thus, pursuant to some embodiments of the present invention, millimeter wave communications systems are provided that include an RF printed circuit board structure. The RF printed circuit board structure may include a phased array antenna that has a plurality of radiating elements and a channel block that includes a plurality of active antenna channels that each feed a respective one of a plurality of sub-groups of the radiating elements. A first sub-set of the active antenna channels may be completely positioned on a first side of the phased array antenna and a second sub-set of the active antenna channels may each include a first portion that is on a second side of the phased array antenna, the second side being adjacent the first side. 
     The second sub-set of the active antenna channels may include, for example, a total of one or two active antenna channels in some embodiments. Second portions of all of the active antenna channels may extend generally in the same direction. The radiating elements may be arranged in a plurality of rows and a plurality of columns, and each active antenna channel may be connected to a respective one of the columns of radiating elements. In some embodiments, each active antenna channel may include a high power amplifier and a low noise amplifier, where the high power amplifier and the low noise amplifier of a first of the active antenna channels is positioned closer to the phased array antenna than are the high power amplifier and the low noise amplifier of the second of the active antenna channels. 
       FIG. 11  is a plan view of an example implementation of one of the power couplers  350 . As shown in  FIG. 11 , the power coupler  350  may be implemented using seven Wilkinson power couplers  354 - 1  through  354 - 7  that are arranged in a root/branch structure so that, for signals travelling in the “transmit” direction, the power coupler  350  has a single input  351  and eight outputs  352  (for signals travelling in the “receive” direction the power coupler  350  has eight inputs  352  and a single output  351 ). Transmission line segments  356  extend between the inputs and outputs of the Wilkinson power dividers  354  and form the outputs  352  of the power coupler  350 . 
     As is further shown in  FIG. 11 , a plurality of meandered delay lines  358  are incorporated into various of the transmission line segments  356 . The meandered delay lines  358  are used to compensate for differences in the path lengths from the input  351  of power coupler  350  to the eight columns  386  of the phased array antenna  380 . If there are delay differences along these eight paths, then the amplitude and/or phase weights that are applied by the variable attenuators  366 ,  378  and variable phase shifters  368 ,  376  in the active antenna channels  362  may only produce a desired beam pattern over a narrower range of frequencies. The meandered delay lines  358  increase the path length on certain of the paths to provide substantially equal path lengths. 
     Each meandered delay line  358  may comprise a transmission line segment that has a series of opposed bends that form a serpentine pattern. In some embodiments, the meandered delay lines  358  may extend along a longitudinal axis and the serpentine pattern may be used to increase the physical length of the transmission path by, for example, a factor of two or three or more. Each meandered delay line  358  may be implemented using a co-planar waveguide transmission line segment, while most or all of the remainder of the transmission lines in power coupler  350  may be implemented as microstrip transmission line segments. Implementing each meandered delay line  358  using a co-planar waveguide transmission line segment may allow for greater transmission line length in a given area, since the co-planar waveguide transmission line segment has a smaller width. Additionally, since the co-planar waveguide transmission line segments have ground conductors on either side of the conductive track, it may do a better job than a microstrip transmission line segment at reducing coupling between adjacent bends of the meandered delay lines  358 . 
     As is also shown in  FIG. 11 , the outputs of Wilkinson power couplers  354 - 1 ,  354 - 2  and  354 - 3  connect to the inputs of Wilkinson power couplers  354 - 4  through  354 - 7 . For example, the first output of Wilkinson power coupler  354 - 1  connects to the input of Wilkinson power coupler  354 - 2  via a first transmission line segment  356 - 1  while the second output of Wilkinson power coupler  354 - 1  connects to the input of Wilkinson power coupler  354 - 3  via a second transmission line segment  356 - 2 . The lengths of transmission line segments  356 - 1  and  356 - 2  are different. The use of transmission line segments having asymmetric lengths in the power coupler  350  allows for equalizing the total path length from the input  351  of power coupler to each of the columns  386  of the phased array antenna  380 . In other words, the asymmetric lengths of the transmission line segments  356  and the meandered delay lines  358  compensate for the differences in path lengths in the active antenna channels  362  that are caused, for example, by having some of the active antenna channels  362  wrap around the phased array antennas  380 . This technique may be used to ensure that all of the sub-components of an RF signal to be transmitted arrive at the first radiating element  382  in each column  386  of the phased array antenna  380  at the same time. 
     Thus, pursuant to some embodiments of the present invention, power couplers for a millimeter wave communications system are provided that include a first 1×2 power coupler having a first input, first output and a second output; a second 1×2 power coupler having a second input that is coupled to the first output by a first transmission line segment; and a third 1×2 power coupler having a third input that is coupled to the second output by a second transmission line segment. The first transmission line segment includes a meandered delay line. The meandered delay line may comprise, for example, a co-planar waveguide meandered delay line and at least another portion of the first transmission line segment may comprise a microstrip transmission line segment. The 1×2 power couplers may be, for example, Wilkinson power couplers. 
     In some embodiments, the power couplers may further include a fourth 1×2 power coupler having a fourth input and a fifth 1×2 power coupler having a fifth input. In such embodiments the second 1×2 power coupler may have a third output that is coupled to the fourth input by a third transmission line and a fourth output that is coupled to the fifth input by a fourth transmission line, where the third transmission line is longer than the fourth transmission line. 
       FIG. 12  is a schematic diagram illustrating heat sink structures  550  that are included in an example implementation of the printed circuit board structure  320  of  FIGS. 5A-5B  to dissipate heat generated in the integrated circuit chips mounted thereon. As shown in  FIG. 12 , each heat sink structure  550  may comprise a dense pattern of conductive vias  552  that are formed through the printed circuit board structure  320 . The conductive vias  552  may be formed directly underneath various of the integrated circuit chips, including the amplifier integrated circuit chips, to provide a thermal conduction path that may be used to vent heat generated in the integrated circuit chips out of the printed circuit board structure  320 . Referring to  FIGS. 4B and 13 , a top end of each conductive via  552  may contact an integrated circuit chip  315 , while a bottom end of each conductive via  552  may contact one of the pedestals  314  of the baseplate  310 . 
       FIG. 13  is a plan view of an example implementation of the printed circuit board structure  320  of  FIGS. 5A-5B  that illustrates via fence structures  700  and DC bias coupling circuits  720  that are implemented in the printed circuit board structure  320 . The via fence structures  700  are provided between the microstrip transmission lines  710  that connect each active antenna channel  362  to a column  386  of one of the phased array antennas  380 . As shown in  FIG. 13 , in regions where two of these microstrip transmission lines  710  extend in parallel in close proximity to each other, a via fence structure  700  may be formed between the adjacent microstrip transmission lines  710 . Each via fence structure  700  may comprise a row of conductive vias  702  that extend through the printed circuit board structure  320 . A metal trace  704  may be provided on the metallization layer  324 - 1  that electrically connects the conductive vias  702  of the via fence structure  700 . The metal trace  704  may improve the isolation between adjacent microstrip transmission lines  710 . 
       FIG. 13  also illustrates DC bias signal injection circuits  720  that may be used to inject DC control signals onto the microstrip transmission lines  710  that are used to control the PIN diode switches that are included in each column  386  of each phased array antenna that are used to implement elevation beamwidth switching. The PIN diode switches and operation thereof are described in greater detail in the aforementioned U.S. Provisional Patent Application Ser. No. 62/506,100, entitled  Phased Array Antennas Having Switched Elevation Beamwidths and Related Methods,  Attorney Docket No. 9833-1221-PR, filed May 15, 2017. The DC bias signal injection circuits  720  may be implemented as a narrow, quarter wavelength long conductive trace  722  that connects a DC bias voltage source (here a conductive via  724  that carries a DC voltage) to one of the microstrip transmission lines  710 . Each third transmit/receive switch  372  includes a capacitor that also acts as part of the respective DC bias signal injection circuits  720 . DC bias signals are passed from the conductive via  724  to the microstrip transmission line  710  through the quarter wavelength long conductive trace  722 . The quarter wavelength long conductive trace  722  blocks RF signals carried on microstrip transmission line  710  from flowing to the conductive via  724 . The capacitor in each third transmit/receive switch  372  prevents the DC bias signal from flowing into the active antenna channels  362 . 
       FIG. 14  is a plan view of an example implementation of the four phased array antennas  380  included in the printed circuit board structure  320 . As can be seen in  FIG. 14 , each patch radiating element  382  in a column  386  connects to a feeder transmission line segment  388  via a feed  389 . In the phased array antennas  380 - 1 ,  380 - 2  that are located at the top of the figure, each feed  389  defines a line that bisects the patch radiating element  382  at an angle of −45 degrees with respect to an axis V while in the phased array antennas  380 - 3 ,  380 - 4  that are located at the bottom of the figure, each feed  389  defines a line that bisects the radiating element at an angle of +45 degrees with respect to the axis V. Thus, the patch radiating elements  382  in phased array antennas  380 - 1 ,  380 - 2  will transmit and receive signals at a −45 degree polarization, while the patch radiating elements  382  in phased array antennas  380 - 3 ,  380 - 4  will transmit and receive signals at a +45 degree polarization. 
     By implementing the phased array antennas  380  to have two different polarizations, increased isolation may be provided between the phased array antennas  380  when the millimeter wave communications system  60  operates using MIMO transmission techniques. In other words, in addition to spatial diversity, the radio unit  100  will have polarization diversity between some of the phased array antennas  380 . Additionally, since the two phased array antennas  380  in the “top row” have the same polarization, the radio unit  100  may alternatively be operated with the phased array antennas  380 - 1 ,  380 - 2  operating as a first, 16 column antenna and the phased array antennas  380 - 3 ,  380 - 4  operated as a second, 16 column antenna. When operated in this fashion, the radio unit  100  may transmit in a 2×MIMO mode using the two 16-column antennas. By combining two of the phased array antennas  380  into a “larger” antenna the azimuth beamwidth may be further narrowed and the antenna gain increased. In rural areas where subscribers are spaced farther apart operation in 2×MIMO mode may be preferred, while 4×MIMO operation may provide better performance in urban areas. 
     The millimeter wave communications systems according to embodiments of the present invention may support high performance levels. As discussed above, each millimeter wave communications system includes four phased array antennas and hence may support 4×MIMO transmissions to provided increased throughput. Additionally, each antenna may be actively scanned in the azimuth plane and beamwidth switching may be provided in the elevation plane to provide relatively high gain antenna beams while still ensuring that the radio unit may provide coverage to all of the users within its coverage area. The millimeter wave communications systems may support very high effective isotropic radiated power (“EIRP”) levels due to the above-discussed high antenna gains and because the millimeter wave communications systems have multi-stage amplification on the RF board. 
     The millimeter wave communications systems according to embodiments of the present invention may also be very compact and relatively inexpensive. The use of elevation beamwidth switching allows the phased array antennas included in the millimeter wave communications systems according to embodiments of the present invention to have most of the capabilities of a full two dimensional active antenna array while only requiring 12.5% of the active transceivers that are necessary for a fully active 8×8 phased array antenna. This reduction in electronic components provides a highly cost-effective implementation and also reduces the size of the millimeter wave communications system. 
     The present invention has been described above with reference to the accompanying drawings. The invention is not limited to the illustrated embodiments; rather, these embodiments are intended to fully and completely disclose the invention to those skilled in this art. In the drawings, like numbers refer to like elements throughout. Thicknesses and dimensions of some elements may not be to scale. 
     Spatially relative terms, such as “under”, “below”, “lower”, “over”, “upper”, “top”, “bottom” and the like, may be used herein for ease of description to describe one element or feature&#39;s relationship to another element(s) or feature(s) as illustrated in the figures. It will be understood that the spatially relative terms are intended to encompass different orientations of the device in use or operation in addition to the orientation depicted in the figures. For example, if the device in the figures is turned over, elements described as “under” or “beneath” other elements or features would then be oriented “over” the other elements or features. Thus, the exemplary term “under” can encompass both an orientation of over and under. The device may be otherwise oriented (rotated 90 degrees or at other orientations) and the spatially relative descriptors used herein interpreted accordingly. 
     Well-known functions or constructions may not be described in detail for brevity and/or clarity. As used herein the expression “and/or” includes any and all combinations of one or more of the associated listed items. 
     It will be understood that, although the terms first, second, etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first element could be termed a second element, and, similarly, a second element could be termed a first element, without departing from the scope of the present invention. 
     It will be understood that the above embodiments may be combined in any way to provide a plurality of additional embodiments.