Patent Publication Number: US-2011063050-A1

Title: Semiconductor integrated circuit

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is based upon and claims the benefit of priority from Japanese Patent Application No. P2009-213986, filed on Sep. 16, 2009; the entire contents of which are incorporated herein by reference. 
     FIELD 
     Embodiments described herein relate generally to a semiconductor integrated circuit to switch the frequency band of an amplifier. 
     BACKGROUND 
     In recent wireless communication systems for mobile terminals and the like, transmission and reception systems have come to use broader frequency bands and multiple frequency bands to support all sorts of communication systems, such as GSM, GPS, and WCDMA. 
     In a wireless communication system using a frequency band that is not very broad, such as several hundreds of megahertz, if the input and the output have broadband properties, intermodulation caused by in-band and out-of-band interferences or the like sometimes cause distortion components within a desired frequency band. 
     Accordingly, in a system of a narrow frequency band, it is desirable to narrow down the frequency band, that is, to switch the frequency band from one to another, in order to reduce the degradation of the distortion characteristics, the noise characteristics, and the like. 
     A wireless communication system includes a resonant circuit. The resonant circuit includes an inductor and a capacitor, and the resonant frequency of the resonant circuit is expressed as 1/{2pv(LC)}, where L is the inductance (induction coefficient) of the inductor, and C is the capacitance of the capacitor. By changing the inductance or the capacitance, the resonant frequency can be varied, that is, the frequency band can be switched from one to another. 
     In general, a widely-used method of varying the resonant frequency is a frequency tuning method using a passive element such as a resistor. 
     At high frequencies, in particular, a method using inductors or capacitors as the loads on an amplifier is used, such as a method 1 of switching the capacitance in accordance with the frequency band, a method 2 of switching the inductance similarly, or a method 3 of switching the inductance with use of the coupling coefficient of the inductors. 
     However, since the method 1 switches the capacitance, obtaining a broad, variable frequency range requires so a large capacitance that it is difficult to get a high Q-factor at low frequencies. Here, the Q-factor refers to a value expressed as Q=1/Rv(L/C), where R is the parasitic resistance of the inductor, L is the inductance of the coil, and C is the capacitance of the capacitor. The higher the Q-factor is, the steeper the frequency characteristics become and the more highly selective the frequency becomes. 
     In the method 2, the value of the inductor itself is changed by use of a switch, so that the inductors have to be provided in a number corresponding to the systems to which the method is applied, and therefore occupy a larger area. In addition, if a broader variable frequency range is desired, larger inductors are required, so that the parasitic resistance becomes larger and the Q-factor is degraded. 
     In the method 3, by switching between the same-phase mode and the differential mode, the inductance is varied within a range from (1−k)L to (1+k)L in accordance with the coupling coefficient k, and therefore the frequency range can be varied in accordance with the inductance. However, the resonant frequency in the same-phase mode, when the Q-factor is so low as (1/R)v{(1−k)L/C}, is higher than the resonant frequency in the differential mode. In particular, at higher frequencies, in comparison to lower frequencies, it is inherently difficult to obtain favorable characteristics, and the Q-factor of the inductor used instead of the load or the degeneration resistor has a great influence on the gain characteristics, maximum output electric power, oscillation amplitude, and the like in an analogue circuit block, such as a low noise amplifier (LNA), a power amplifier (PA), or a voltage-controlled oscillator (VCO). For those reasons, an element that has as high a Q-factor as possible is preferably used, but the method 3 has a problem that, at higher frequencies, the Q-factor is degenerated and thus desirable gain and noise characteristics cannot be achieved. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a system block diagram of a semiconductor integrated circuit of the invention. 
         FIG. 2  is a circuit diagram of Embodiment 1. 
         FIG. 3  is a circuit diagram of Embodiment 2. 
         FIG. 4  is a circuit diagram of Embodiment 3. 
         FIG. 5  is a circuit diagram of Embodiment 4. 
         FIG. 6  is a circuit diagram of Embodiment 5. 
         FIG. 7  is a circuit diagram of Embodiment 6. 
         FIG. 8  is a circuit diagram of Embodiment 7. 
         FIG. 9  is a circuit diagram of Embodiment 8. 
     
    
    
     DETAILED DESCRIPTION 
     In general, according to one embodiment, a semiconductor integrated circuit including: a first coil and a second coil having a first coupling coefficient and being connected in parallel to each other; a third coil connected in series to the first coil and the second coil; a first capacitor connected in parallel to an end of the first coil and to an end of the third coil; a second capacitor connected in parallel to an end of the second coil and to the end of the third coil; a first input terminal connected to the end of the first coil and to an end of the first capacitor; a second input terminal connected to the end of the second coil and to an end of the second capacitor; and an input-signal supplying portion configured to supply input signals of opposite phases to the first input terminal and the second input terminal, respectively. 
     Some embodiments of the invention will be described below by referring to the drawings. 
     Embodiment 1 
       FIG. 1  is a system block diagram of a semiconductor integrated circuit of the invention. The semiconductor integrated circuit of the invention includes an antenna  10  configured to transmit and receive data, a T/R switching portion  11  configured to switch the transmission and the reception of data, a low noise amplifier (LNA)  12 , a power amplifier (PA)  13 , a mixer (MIX)  14 , a voltage controlled oscillator (VCO)  15 , and a low-pass filter (LPF)  16 , and what is characteristic of the semiconductor integrated circuit of the invention is in the LNA  12  and the PA  13 , in particular. 
       FIGS. 2A and 2B  each show a resonant circuit portion  100  of the semiconductor integrated circuit according to Embodiment 1 of the invention, and the resonant circuit portion  100  is included in the LNA  12  (or in the PA  13 ) shown in  FIG. 1 . The resonant circuit portion  100  shown in  FIG. 2A  includes: a coil L 11  with an inductance L 1  and a coil L 12  with an inductance L 1  which have a coupling coefficient k 1  and are connected in parallel to each other; a coil L 13  with an inductance L 2  connected in series to both the coil L 11  and the coil L 12 ; a capacitor C 11  with a capacitance C 1  connected in parallel with the coil L 11 ; and a capacitor C 12  with a capacitance C 1  connected in parallel with the coil L 12 . 
     An input terminal p 1  is connected both to one end of the coil L 11  and to one end of the capacitor C 11 , and an input terminal n 1  is connected both to one end of the coil L 12  and to one end of the capacitor C 12 . Signals are supplied both to the input terminal p 1  and to the input terminal n 1  from an input-signal supplying portion  200 . 
     The coil L 11  and the coil L 12  are wound in such directions that their respective magnetic fluxes reinforce each other when signals of opposite phases to each other pass through the coil L 11  and the coil L 12 , respectively. 
     When the input-signal supplying portion  200  supplies signals of opposite phases to the input terminal p 1  and to the input terminal n 1 , respectively, signals of opposite phases respectively pass through the coil L 11  and the coil L 12  that form the inductor, so that a magnetic force acts to make the magnetic fluxes reinforce each other. Accordingly, the inductance L L  of the inductor including the coils L 11 , L 12 , and L 13  is expressed as L L =L 1 (1+k 1 ), where k 1  is the coupling coefficient of the coils L 11  and L 12 . In addition, since the signal that passes through the coil L 12  has the opposite phase, the influence of the coil L 13  is negligible. Accordingly, the resonant frequency f L =1/R{(2pv(L L ×C 1 ))}, and the Q-factor Q L =(1/R)v(L L /C 1 ), so that the resonant circuit portion  100  can be made to resonate at low frequencies without degrading the Q-factor. Note that R in the equations above is the parasitic resistance of the corresponding inductor (R means the same in the following descriptions). 
     The resonant circuit portion  100  shown in  FIG. 2B  is the same as the one shown in  FIG. 2A  except the phase of the signal to be supplied to the input terminal n 1 . In  FIG. 2B , signals of the same phase are supplied to the input terminal n 2  and the input terminal p 2  from the input-signal supplying portion  200 . When signals of the same phase are supplied to the input terminals p 2  and n 2 , signals of the same phase respectively pass through the coil L 21  and the coil L 22  that form the inductor, so that a magnetic force acts to make the magnetic fluxes cancel each other out. Accordingly, the inductance of the inductor including the coils L 21  and L 22  is expressed by L 1 (1−k 1 ), where k 1  is the coupling coefficient of the coils L 21  and L 22 . In addition, since the signal that passes through the coil L 21  is a signal of the same phase, the inductance L H  of the inductor including the coils L 21 , L 22 , and L 23  is expressed as L H =2L 2 +L 1 (1−k 1 ). In addition, the resonant frequency f H =1/{2pv(L H C 1 )} and the Q-factor Q H =(1/R)v(L H /C 1 ). Accordingly, while sufficient isolation is secured, the frequency can be set to a desired one. In addition, because the Q-factor in the high-frequency operation mode can be improved, favorable gain characteristics and noise characteristics can be obtained. Specifically, while the Q factor of the coils L 11  and L 12  and the Q-factor of the coils L 21  and L 22  are degraded by the same-phase mode, each of the degraded Q-factors can be compensated by the coil L 13 /L 23 , so that a high Q-factor can be secured even in the high-frequency mode, and favorable gain characteristics and noise characteristics can be obtained. 
     Embodiment 2 
       FIG. 3  shows a resonant circuit portion  100  of a semiconductor integrated circuit according to Embodiment 2 of the invention. The resonant circuit portion  100  shown in  FIG. 3A  includes: a coil L 11  with an inductance L 1  and a coil L 12  with an inductance L 1  which have a coupling coefficient k 1  and are connected in parallel to each other; a coil L 21  with an inductance L 1  and a coil L 22  with an inductance L 1  which have also a coupling coefficient k 1  and are connected in parallel to each other; a coil L 13  with an inductance L 2  which is connected in series both to the coil L 11  and to the coil L 12 ; a coil L 23  with an inductance L 2 , which is connected in series both to the coil L 21  and to the coil L 22 ; a capacitor C 11  with a capacitance C 1  which is connected in parallel to the coil L 11 ; a capacitor C 12  with a capacitance C 1  which is connected in parallel to the coil L 12 ; a capacitor C 21  with a capacitance C 1  which is connected in parallel to the coil L 21 ; and a capacitor C 22  with a capacitance C 1  which is connected in parallel to the coil L 22 . 
     Note that the coil L 13  and the coil L 23  have a coupling coefficient k 2 . 
     The coil L 11  and the coil L 12  are wound in such directions that their respective magnetic fluxes reinforce each other when signals of opposite phases pass through the coils L 11  and L 12 , respectively. In addition, the coil L 21  and the coil L 22  as well as the coil L 13  and the coil L 23  are wound in the same manner. 
     An input terminal p 1  is connected both to one end of the coil L 11  and to one end of the capacitor C 11 , and an input terminal n 1  is connected both to one end of the coil L 12  and to one end of the capacitor C 12 . Moreover, an input terminal p 2  is connected both to one end of the coil L 21  and to one end of the capacitor C 21 , and an input terminal n 2  is connected both to one end of the coil L 22  and to one end of the capacitor C 22 . Signals are supplied to all the input terminals p 1 , n 1 , p 2 , and n 2  from an input-signal supplying portion  200 . 
     When signals of opposite phases are supplied respectively to the input terminal p 1  and to the input terminal n 1 , signals of opposite phases respectively pass through the coil L 11  and the coil L 12  forming the inductor, so that a magnetic force acts to make the magnetic fluxes reinforce each other. Accordingly, the inductance L L  of the inductor including the coils L 11 , L 12 , and L 13  is expressed as L L =L 1 (1+k 1 ), where k 1  is the coupling coefficient of the coils L 11  and L 12 . Likewise, the inductance L L  of the inductor including the coils L 21  and L 22  is expressed as L L =L 1 (1+k 1 ), where k 1  is the coupling coefficient of the coils L 21  and L 22 . 
     In addition, since the signal that passes through each of the coils L 12  and L 22  has the opposite phase, the influence of each of the coils L 13  and L 23  is negligible. Accordingly, the resonant frequency f L =1/R{(2pv(L L ×C 1 ))}, and the Q-factor Q L =(1/R)v(L 1 /C 1 ), so that the resonant circuit portion  100  can be made to resonate at low frequencies without degrading the Q-factor. 
     The resonant circuit portion  100  shown in  FIG. 3B  is the same as the one shown in  FIG. 3A  except the phases of the signals to be supplied to the input terminal n 1  and p 2 . In  FIG. 3B , signals of the same phase are supplied to the input terminals p 1  and n 1 , and signals of the same phase are supplied to the input terminals p 2  and n 2 . Signals to be input into the respective input terminals p 1  and p 2  have opposite phases, and signals to be input into the respective input terminals n 1  and n 2  have opposite phases. 
     When signals of the same phase are supplied to the input terminals p 1  and n 1 , signals of the same phase pass through the coil L 11  and the coil L 12 , so that a magnetic force acts to make the magnetic fluxes cancel each other out. In addition, when signals of the same phase are supplied to the input terminals p 2  and n 2 , signals of the same phase pass through the coil L 21  and the coil L 22 , so that a magnetic force acts to make the magnetic fluxes cancel each other out. Accordingly, the inductance of the inductor including the coils L 11  and L 12  is expressed by L 1 (1−k 1 ), where k 1  is the coupling coefficient of the coils L 11  and L 12 . Likewise, the inductance of the inductor including the coils L 21  and L 22  is expressed by L 1 (1−k 1 ), where k 1  is the coupling coefficient of the coils L 21  and L 22 . 
     In addition, since signals of opposite phases respectively pass through the coil L 13  and the coil L 23 , so that a magnetic force acts to make the magnetic fluxes reinforce each other. Accordingly, the inductance of the inductor including the coils L 13  and L 23  is expressed by L 2 (1+k 2 ), where k 2  is the coupling coefficient of the coils L 13  and L 23 . 
     Accordingly, the inductance L H  of the inductor including the coils L 11 , L 12 , and L 13  is expressed as L H =2L 2 (1+k 2 )+L 1 (1−k 1 ). In addition, the resonant frequency f H =1/{2pv(L H C)}, and the Q-factor Q H =(1/R)v(L H /C), so that the resonant circuit portion  100  can be made to resonate at high frequencies without degrading the Q-factor. 
     Note that, if the coils L 13  and L 23  having a coupling coefficient k 2  are not provided, the inductance L H ′ of such a case is expressed as L 1 (1−k 1 ), which means that the Q-factor of this Embodiment 2 is v(L H /L H ′) times higher than the Q-factor of this case. Accordingly, the resonant circuit portion  100  can be made to resonate at high frequencies without degrading the Q-factor. 
     Embodiment 3 
       FIG. 4  shows a semiconductor integrated circuit according to Embodiment 3 of the invention. Embodiment 3 is a differential amplifier circuit of a cascade configuration including the resonant circuit portion  100  of Embodiment 2 and an input-signal supplying portion  200  including switches M 21  to M 26  configured to selectively switch signals to supply either signals of the same phase or signals of opposite phases to the coils L 11  and L 12 , respectively, and to the coils L 21  and L 22 , respectively. 
     To resonate the resonant circuit portion  100  at low frequencies, signals of opposite phases are supplied to the resonator by turning the switches M 21 , M 23 , M 24 , and M 26  ON (H), and by turning the switches M 22  and M 25  OFF (L). 
     As in Embodiment 2, the inductance L L , the resonant frequency f L , and the Q-factor Q L  are expressed as L L =L 1  (1+k 1 ), f L =1/{2pv(L L C)}, and Q L =(1/R)v(L L /C), respectively. 
     To resonate the resonant circuit portion  100  at high frequencies, signals of the same phase are supplied to the resonator by turning the switches M 21 , M 22 , M 24 , and M 25  ON (H), and by turning the switches M 23  and M 26  OFF (L). 
     As in Embodiment 2, the inductance L H , the resonant frequency f H , and the Q-factor Q H  are expressed as L H =2L 2 (1+k 2 )+L 1 (1−k 1 ), f H =1/{2pv(L H C)}, and Q H =(1/R)v(L H /C), respectively. 
     Accordingly, besides the effects obtained by Embodiment 2, Embodiment 3 can switch selectively the signals to supply either signals of the same phase or signals of opposite phases to the respective coils. 
     Embodiment 4 
       FIG. 5  shows a circuit of Embodiment 4 of the invention. In the circuit of Embodiment 4, an LC resonant circuit is provided in the input-signal supplying portion  200  of Embodiment 3 by further providing a variable capacitance capacitor C 31  in parallel to the coil L 31  and a variable capacitance capacitor C 32  in parallel to the coil L 32 . 
     By changing the capacitance of the switched-capacitor C 31  and the capacitance of the switched-capacitor C 32 , not only the effects obtained in Embodiments 1 to 3 but also an additional effect can be obtained. Specifically, the operation mode can be switched, when necessary, between the high-frequency operation mode and the low-frequency operation mode. Note that the impedance of the LC resonant circuit is made sufficiently high at desired operation frequencies. 
     Embodiment 5 
       FIG. 6  shows a circuit of Embodiment 5 of the invention. The circuit of Embodiment 5 is a complementary cascade-connected gate-grounded differential amplifier circuit including: two resonant circuits each being the resonant circuit  100  of Embodiment 2; and an input-signal supplying portion  200  including switches MN 21  to MN 26  and MP 21  to MP 26  configured to selectively switch signals to supply either signals of the same phase or signals of opposite phases to the coils L 11  and L 12 , respectively, and to the coils L 21  and L 22 , respectively. The resonant circuit portion  100  of Embodiment 5 is controlled in the same way as in the case of Embodiment 3, and the control signals to control the switches MP 21  to MP 26  are the reverse signals of the control signals for the switches MN 21  to MN 26 . An output signal VOUT_P 1  and an output signal VOUT_P 2  are signals of the same phase, so that these output signals are output by summing the currents. 
     Embodiment 6 
       FIG. 7  shows a circuit of Embodiment 6 of the invention. The circuit of Embodiment 6 is an amplifier circuit including: a resonant circuit portion  100  that is similar to the one in Embodiment 2; and an input-signal supplying portion  200  including switches M 21  to M 26  and M 31  and M 32  configured to selectively switch signals to supply either signals of the same phase or signals of opposite phases to the coils L 11  and L 12 , respectively, and to the coils L 21  and L 22 , respectively. While the switches M 11  and M 12  operate as a common-source amplifier, the switches M 31  and M 32  operate as a common-gate amplifier. The common-source circuit and the common-gate circuit are an inverting amplifier and a noninverting amplifier, respectively, so that, if the transconductance of the switches M 11  and M 12 , and the transconductance of the switches M 31  and M 32  are made to be the same transconductance Gm, signals with the same amplitude and opposite phases pass through the resonant circuit portion  100 . In addition, DC current sources I 1  and I 2  are additionally provided as the DC current sources for the switches M 31  and M 32 . 
     To resonate the resonant circuit portion  100  at low frequencies, signals of opposite phases are supplied to the resonator by turning the switches M 21 /M 22 , and M 24 /M 25  ON (H), and by turning the switches M 23 /M 26  OFF (L). Signals of opposite phases pass respectively through the inductor including the coils L 11  and L 12  and through the inductor including the coils L 21  and L 22 , so that a magnetic force acts to make the magnetic fluxes reinforce each other. Accordingly the resonant circuit portion  100  resonates at low frequencies. 
     To resonate the resonant circuit portion  100  at high frequencies, signals of the same phase are supplied to the resonator by turning the switches M 21 /M 23 , and M 24 /M 26  ON (H), and by turning the switches M 22 /M 25  OFF (L). Signals of the same phase respectively pass through the inductor including the coils L 11 /L 12  and the inductor including the coils L 21 /L 22  so that a magnetic force acts to make the magnetic fluxes cancel each other out. Accordingly the resonant circuit portion  100  resonates at high frequencies. 
     Embodiment 7 
       FIG. 8  shows a circuit of Embodiment 7 of the invention. The circuit of Embodiment 7 is an amplifier circuit including: a resonant circuit portion  100  that is similar to the one in Embodiment 2; and switches M 21  to M 26  configured to selectively switch signals to supply either signals of the same phase or signals of opposite phases to the coils L 11  and L 12 , respectively, and to the coils L 21  and L 22 , respectively. In addition, the switches M 11 /M 12  operate as a common-drain amplifier circuit. The LC resonator is controlled in the same way as in the case of Embodiment 1, and is used mainly as a buffer circuit. 
     Embodiment 8 
       FIG. 9  shows a circuit of Embodiment 8 of the invention. The circuit of Embodiment 8 includes a resonant circuit portion  100  that is similar to the one in Embodiment 2 and an input-signal supplying portion  200  including: current sources I 2 /I 3  and buffer elements M 21  to M 24  configured to selectively switch signals to supply either signals of the same phase or signals of opposite phases to the coils L 11 /L 12 , respectively, and to the coils L 21 /L 22 , respectively. In addition, a current source I 1  is provided to supply current to M 11 /M 12 . 
     The voltage signals input into the gates of M 21  to M 24  appear as inverted signals at the drains of their respective transistors. Accordingly, by turning ON/OFF the current sources I 2  and I 3  selectively, the signal input into the gates of M 22  and M 23 , and the signal input into the gates of M 21  and M 24  can switch the voltage signals at the drains of M 11  and M 12  to have either the same phase or opposite phases. Accordingly, the circuit of Embodiment 8 can change the frequency of the output signal as in the case of Embodiment 2. 
     Other Embodiments 
     The capacitors C 11 , C 12 , C 21  and C 22  in Embodiments 1 to 7 may be replaced with variable capacitors (varactors). By this replacement, the frequency can be changed, so that the Q-factor of the coils L 11  and L 12  and that of the coils L 21  and L 22  degraded in the same-phase mode can be compensated by the coils L 13  and L 23  as well as the variable capacitors C 11 , C 12 , C 21  and C 22 . Thus, while a broad variable frequency range is secured, the frequency band can be tuned finely. In addition, a high Q-factor can be secured even in the high-frequency mode, so that favorable gain characteristics and noise characteristics can be obtained. 
     If, in each Embodiment, the inductor including the coils L 11 /L 12  and the inductor including the coils L 21 /L 22  are used with the same phase, the output signals may be summed together to make the signal level higher than in the case of the output from either one alone. 
     In addition, the amplifier including the resonant circuit portion  100  can be formed not only as an FET but also as a bipolar one. Besides, the amplifier can be formed to be a PMOS one with reversed polarity. 
     While certain embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the inventions. Indeed, the novel devices and methods described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the embodiments described herein may be made without departing from the spirit of the inventions. The accompanying claims and their equivalents are intended to cover such forms or modification as would fall within the scope and spirit of the inventions.