Patent Publication Number: US-2022240024-A1

Title: Hearing device and method of operating the hearing device

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims the priority, under 35 U.S.C. § 119, of German Patent Application DE 10 2021 200 642.6, filed Jan. 25, 2021; the prior application is herewith incorporated by reference in its entirety. 
     FIELD AND BACKGROUND OF THE INVENTION 
     The invention relates to a hearing device, in particular hearing aid, containing a transmitter circuit for wireless signal transmission. The invention furthermore relates to a method for operating such a hearing device. 
     Hearing aids are portable hearing devices (hearing instrument) that are used to provide support for persons who are hard of hearing or have a hearing impediment. In order to meet the numerous individual needs, different designs of hearing aids are provided, such as behind-the-ear hearing devices (BTE) and hearing devices with an external receiver (RIC: receiver in the canal) and also in-the-ear hearing devices (ITE), including for example concha hearing devices or channel hearing devices (ITE: In-The-Ear, CIC: Completely-In-Channel, IIC: Invisible-In-The-Channel). The hearing devices listed as examples are worn on the external ear or in the auditory canal of a hearing aid user. Furthermore, however, bone conduction hearing aids, implantable hearing aids or vibrotactile hearing aids are also available on the market. In these devices, impaired hearing is stimulated either mechanically or electrically. 
     In principle, such hearing devices have an input transducer, an amplifier and an output transducer as essential components. The input transducer is generally an acousto-electric transducer, such as a microphone, for example, and/or an electromagnetic receiver, for example an induction coil or a (radio-frequency, RF) antenna. The output transducer is usually realized as an electro-acoustic transducer, for example as a miniature loudspeaker (receiver), or as an electromechanical transducer, such as a bone conduction receiver, for example. The amplifier is usually integrated into a signal processing device. Energy is usually supplied by a battery or a rechargeable battery. 
     In the case of a so-called binaural hearing aid, two such hearing devices are worn by a user, there being a communication or signal connection between the hearing devices, this also being referred to as ear-to-ear (e2e) communication. In this case, during operation, data, possibly even large amounts of data, and/or audio signals are exchanged, for example wirelessly, between the hearing devices on the right and left ears. The data and information exchanged enable the hearing devices to be adapted particularly effectively to a respective acoustic environmental situation. In particular, this enables particularly authentic spatial acoustics for the user and improves the intelligibility of speech, even in noisy environments. Furthermore, features and functionalities such as narrow focus or CROS (Contralateral Routing Of Signal) are made possible, for example. 
     On account of the limited energy sources in a hearing device, it is necessary for the e2e signal transmission to be embodied in a manner that saves as much power as possible, in order to enable a battery life of several days with active e2e communication. In this case, e2e communication systems are typically realized as a magneto-inductive connection consisting of an analog and digital transmitter, two coils as transmitting and receiving antenna and an analog and digital receiver. A transmitter circuit of the communication system is critical here with regard to battery consumption. 
     In the course of signal transmission, a payload signal to be transmitted is regularly altered (modulated) with a so-called carrier, thereby enabling a low-frequency payload signal to be transmitted at high frequency. The payload signal or the data and information are recovered at the reception end by means of a corresponding demodulation. A constant envelope phase modulation technique is generally used as modulation format in the case of binaural hearing devices for energy saving reasons. Other transmitter topologies, such as e.g. IQ modulators, are not suitable for low-power hearing devices on account of the high power consumption. 
     The challenge in respect of design is to construct a power-saving transmitter which generates a signal with constant envelope phase modulation. On account of the limited structural space in the hearing device, it is necessary for the transmitter to be embodied as compactly as possible and in a manner allowing it to be integrated into a customized ASIC. In particular, the intention is to use the fewest possible external components. Furthermore, the transmission signal generated must comply with regulatory specifications (e.g. spectral masks). 
     Such a transmitter or transmitter circuit comprises a resonant circuit and an energy feed-in circuit, for example. 
     The different phases of the transmitter (TX) are realized by the transmission frequency being varied by means of a detuning of the resonant circuit for a specific time duration until a desired phase is attained. In this case, at least three different frequencies are used, firstly the nominal transmission frequency (f 0 ), and secondly a frequency below the transmission frequency (fm) which is used for negative phase shifts, and also a frequency above the transmission frequency (fp) which is used for positive phase shifts. The transmission signal is a continuous signal with a constant envelope. 
     The energy feed-in circuit is embodied as an H-bridge circuit, for example. The energy feed-in circuit compensates for losses and feeds energy into the resonant circuit at an exact point in time in order to maintain the oscillation with constant amplitude and the desired frequency. In this case, the (feed-in) point in time is effected in a manner synchronized with the oscillation. 
     SUMMARY OF THE INVENTION 
     The invention is based on the object of specifying a particularly suitable hearing device. In particular, the intention is to realize particularly energy-saving and, in terms of structural space, compact e2e communication. The invention is furthermore based on the object of specifying a particularly suitable method for operating such a hearing device. 
     The object is achieved according to the invention by means of the features of the independent hearing device claim and by means of the features of the independent method claim. The dependent claims relate to advantageous configurations and developments. The advantages and configurations mentioned in relation to the hearing device are analogously also applicable to the method, and vice versa. 
     The hearing device is designed in particular as a hearing aid, and is preferably used to provide support for a user having a hearing impediment (hearing device user). In this case, the hearing device is designed to pick up sound signals from the environment and to output them to a user of the hearing device. For this purpose, the hearing device contains at least one acousto-electric input transducer, in particular a microphone, and also at least one electro-acoustic output transducer, for example a receiver. During operation of the hearing device, the input transducer picks up sound signals (noises, sounds, speech, etc.) from the environment and converts them into an electrical input signal (acoustic data). An electrical output signal is generated from the electrical input signal by the input signal being modified in a signal processing facility. The signal processing facility is part of the hearing device, for example. The input transducer and the output transducer and optionally the signal processing facility as well are accommodated in particular in a housing of the hearing device. The housing is designed in such a way that it can be worn by the user on the head and near the ear, e.g. in the ear, on the ear or behind the ear. The hearing device is preferably designed as a BTE hearing device, an ITO hearing device or an RIC hearing device. 
     The hearing device comprises a transmitter having a transmitter circuit for wireless signal transmission, in particular for e2e communication. The transmitter circuit contains an electrical resonant circuit having at least one controllable semiconductor switch and having at least one capacitor and also having a transmitter coil or transmission coil. According to the invention, the at least one semiconductor switch is driven by a pulse phase modulator. In other words, the transmitter circuit according to the invention is provided for a pulse phase modulation of an envelope of a transmission signal, and is also suitable and set up therefor. A particularly suitable hearing device is realized as a result. 
     In one suitable development, the resonant circuit comprises two capacitors and two controllable semiconductor switches, wherein a respective semiconductor switch is assigned to a capacitor, and wherein the transmitter coil is interconnected between the capacitors. As a result, the transmitter coil is driveable symmetrically via the capacitors and semiconductor switches. 
     In one advantageous embodiment, an output side of a bridge circuit is connected to the transmitter coil. In other words, the transmitter coil is interconnected between two bridge branches. In this case, the bridge circuit acts as a feed-in circuit provided for feeding loss energy (on account of parasitic losses or radiated power) into the resonant circuit in order to ensure a constant amplitude in the resonant circuit or of the transmission signal. The bridge circuit is preferably embodied as an H-bridge circuit. 
     In one expedient configuration, the pulse phase modulator and the bridge circuit are controlled with the aid of a common timer. In other words, the operating states of the bridge circuit and of the pulse phase modulator are derived from a common timer or clock generator. As a result, it is possible to suitably switch the pulse phase modulator and the bridge circuit while the resonant circuit operates in resonance at a desired transmission frequency, and without the monitoring of an actuating voltage in the capacitor or a current in the transmitter coil. 
     One additional or further aspect of the invention provides for the hearing device to be of binaural design and for this purpose to comprise two individual devices, which each comprise at least one input transducer and also at least one output transducer and are thereby designed to pick up sound signals from the environment and to output them to a user of the hearing device. In addition, each of the individual devices contains a transmitter circuit as a wireless interface for data exchange between the two individual devices. In this case, the individual devices are coupled or at least couplable to one another in terms of signaling via the transmitter circuits. 
     In the case of a binaural hearing device, the two individual devices are worn by the user on different sides of the head, such that each individual device is assigned to an ear. As an alternative to a binaural hearing device, however, a monaural hearing device having just one individual device is also suitable. The explanations regarding a monaural hearing device are analogously applicable to a binaural hearing device, and vice versa. 
     The method according to the invention is provided for operating a hearing device described above, and is also suitable and configured therefor. For wireless signal transmission a transmission signal (TX signal) is generated by the transmitter circuit, wherein according to the method the transmitter coil is disconnected from the or each capacitor at a first point in time, and wherein the transmitter coil is connected to the or each capacitor again at a later second point in time when a desired phase angle of the resonant circuit is attained. In other words, according to the method, the resonant circuit is halted or stopped at the first point in time and is started again at the second point in time, such that a phase offset of the resulting transmission signal is effected. It is thereby possible to switch over the transmitter circuit between different phases (TX phases). 
     The transmitter concept according to the invention thus modulates the phase of the transmission signal by the resonant circuit being paused for a time duration. The method has the advantage that a desired phase is attained almost immediately, but at least within a transmission cycle. In contrast thereto, a phase shift by means of frequency detuning of the resonant circuit in accordance with the prior art, depending on the desired phase step and the frequencies (fm, fp), requires a plurality of transmission cycles until a desired TX phase is attained. 
     Furthermore, a particularly cost-effective pulse phase modulator having a reduced complexity is thus made possible since only one frequency—the nominal transmission frequency (f 0 )—is transmitted. Further transmission frequencies (fm, fp) are no longer required according to the invention, as a result of which the transmitter circuit is simplified. Furthermore, a need for trimming these frequencies is obviated. 
     Insofar as method steps are described below, advantageous configurations for the hearing device arise in particular by virtue of said hearing device being designed to carry out one or more of said method steps. In particular, the hearing device preferably contains a controller (that is to say a control unit) coupled to the transmitter circuit. In this case, the controller can be part of a signal processing facility of the hearing device, for example. 
     In this case, the controller is configured generally—in terms of programming and/or circuitry—for carrying out the above-described method according to the invention. The controller is thus specifically configured to drive the pulse phase modulator and/or the semiconductor switches and also optionally the bridge circuit. 
     In one preferred configuration, the controller is at least essentially formed by a microcontroller with a processor and a data memory, in which the functionality for carrying out the method according to the invention is implemented in terms of programming in the form of operating software (firmware), such that the method—if appropriate in interaction with a device user —is carried out automatically upon the execution of the operating software in the microcontroller. Alternatively, however, in the context of the invention, the controller can also be formed by a non-programmable electronic component, such as an application-specific integrated circuit (ASIC), for example, in which the functionality for carrying out the method according to the invention is implemented with circuitry means. 
     In one advantageous development, the transmitter coil is disconnected from the or each capacitor when the or each capacitor is substantially fully charged. For a good power efficiency, the resonant circuit is stopped in particular when the current in the transmitter coil is zero and the charge in the capacitor has reached a positive maximum by virtue of the transmitter coil being disconnected from the capacitor. 
     Voltage spikes or unwanted artefacts in the transmission signal can occur on account of asymmetries in the transmitter circuit or a detuned resonant circuit. In order to avoid or suppress such unwanted effects, one expedient embodiment provides for the transmitter coil to be short-circuited when the transmitter coil is disconnected from the or each capacitor. The remaining energy in the transmitter coil is short-circuited as a result. 
     With the method according to the invention, a continuous transmission signal is not generated since the resonant circuit is stopped in the case of phase shifts. This leads to signal emissions outside a useful frequency band, too, which are typically higher than during continuous operation. In order to reduce these undesired emissions, one preferred embodiment of the method provides for the or each semiconductor switch to be driven with a control signal of the pulse phase modulator, wherein the control signal contains dither noise. In other words, provision is made of dithering for the driving of the semiconductor switches. 
     By way of example, the semiconductor switches are opened when the assigned capacitor has attained a positive or negative voltage maximum. As a result, two switching processes are realized during a sinusoidal transmission signal (sine wave) at 0° and 180°. For the restart, for a given TX phase it is therefore possible, for example, to choose the second point in time such that the TX phase starts with a positive half-cycle, or that the TX phase starts one half-cycle earlier in this regard, i.e. with a 180° earlier negative half-cycle, or that the TX phase starts one half-cycle later in this regard, i.e. with a 180° later negative half-cycle. In this case, the point in time of switching of the semiconductor switches is preferably changed randomly between these points in time (dithering). This has the advantage that undesired transmissions are significantly reduced and the length of the transmitted TX phase is equalized. 
     During operation, the pulse phase modulator generates, in the transmission signal generated, first-order sidelobes at f 0 ±1.5 fs, wherein f 0  is the carrier frequency and fs is the modulation symbol rate. If the symbol rate is chosen to be very high in order to increase the data rate of the radio link, it can happen that the sidelobes no longer lie within the permissible legal bandwidth limits, as a result of which they have to be damped to a sidelobe level that is predefined by an emission mask. 
     For wireless signal transmission in the case of a hearing device, reducing the sidelobes as simply as possible is desired in order to minimize the processing complexity and thus the power consumption. In one preferred configuration of the method, provision is made for the symbol rate of the transmission signal to be switched over between at least two symbol rate values. In other words, the symbol rate is not constant, but rather changes back and forth between two or more symbol rates. The transmission signal thus results from a superposition of the different symbol rates, as a result of which the sidelobes are suppressed. In this case, the switchover or changeover sequence between the symbol rates is stored for the signal transmission or e2e communication both at the transmitter end and at the receiver end. 
     Other features which are considered as characteristic for the invention are set forth in the appended claims. 
     Although the invention is illustrated and described herein as embodied in a hearing device, it is nevertheless not intended to be limited to the details shown, since various modifications and structural changes may be made therein without departing from the spirit of the invention and within the scope and range of equivalents of the claims. 
     The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         FIG. 1  is a schematic illustration showing a binaural hearing device; 
         FIG. 2  is a block diagram showing a simplified resonant circuit of a transmitter circuit; 
         FIG. 3  is a block diagram showing a transmitter circuit; 
         FIG. 4  is an illustration showing a pulse phase modulation of a transmission signal of the transmitter circuit in one time-coil current diagram formed from five switching pulse diagrams; 
         FIG. 5  shows switching dithering in two time-coil current diagrams; 
         FIG. 6  is a frequency-amplitude diagram showing a spectrum of the transmission signal; 
         FIG. 7  is a frequency-amplitude diagram showing a spectrum for two different symbol rates; and 
         FIG. 8  is a frequency-amplitude diagram showing a spectrum of the transmission signal resulting from the two superposed symbol rates. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Mutually corresponding parts and variables are always provided with the same reference signs in all of the figures. 
     Referring now to the figures of the drawings in detail and first, particularly to  FIG. 1  thereof, there is shown the basic construction of a hearing device  2  according to the invention. In this exemplary embodiment, the hearing device  2  is embodied as a binaural hearing aid containing two hearing aid devices or individual devices  4   a ,  4   b  coupled in terms of signaling. In this case, the individual devices  4   a ,  4   b  are configured for example as behind-the-ear hearing aid devices (BTE). The individual devices  4   a ,  4   b  are coupled or couplable to each other in terms of signaling by means of a wireless signal connection or e2e communication  6 . 
     The construction of the individual devices  4   a ,  4   b  is explained below by way of example on the basis of the individual device  4   a . As illustrated schematically in  FIG. 1 , the individual device  4   a  contains a device housing  8 , in which one or more microphones, also referred to as acousto-electric input transducers  10 , are installed. The input transducers  10  pick up a sound or the acoustic signals in an environment of the hearing device  2  and convert same into an electrical audio signal or acoustic data. 
     The acoustic data are transmitted via lines  12  to a signal processing facility  14 , which is likewise arranged in the device housing  10  and which processes the acoustic data. On the basis of the audio signal, the signal processing facility  14  generates an output signal, which is routed to a loudspeaker or receiver  18  via a line  16 . In this case, the receiver  18  is embodied as an electro-acoustic output transducer, which converts the electrical output signal into an acoustic signal. In the case of the BTE individual device  4   a , the acoustic signal is transmitted to the eardrum of a hearing device user optionally via a sound tube or external receiver, not illustrated in more specific detail, which is connected to an earmold fitted in the auditory canal. However, an electromechanical output transducer as receiver  18  is also conceivable, for example, as in the case of a bone conduction receiver, for example. 
     Energy is supplied to the individual device  4   a  and in particular to the signal processing facility  14  by means of a battery  20  accommodated in the device housing  8 . 
     The signal connection  6  is embodied for example as a magneto-inductive couple between the individual devices  4   a  and  4   b . For this purpose, the signal processing facility  14  is linked to a transmitter  22  in terms of signaling. The transmitter  22  serves for transmitting wireless signals by means of the signal connection  6 . 
     The transmitter  22  contains a transmitter circuit  24  for generating a transmission signal  26  communicated via the signal connection  6 , which transmitter circuit will be explained in greater detail below with reference to  FIGS. 2 and 3 . 
       FIG. 2  shows a schematically simplified illustration of an electrical resonant circuit  28  of the transmitter circuit  24 . The resonant circuit  28  is embodied in particular as a parallel resonant circuit and in this case contains a transmitter coil or transmission coil  30  and a capacitator  32  and also a semiconductor switch  34  embodied as a transistor, in particular as a MOSFET (metal oxide semiconductor field effect transistor). In this case, the semiconductor switch  34  is driven by a pulse phase modulator  36 . 
       FIG. 3  shows the transmitter circuit  24 . In this case, the resonant circuit  28  contains two capacitors  32   a ,  32   b  and two semiconductor switches  34   a ,  34   b . In this case, the terminals of the transmitter coil  30  are contacted via a respective capacitor  32   a ,  32   b , wherein the capacitors  32   a ,  32   b  are respectively assigned one of the semiconductor switches  34   a ,  34   b . The semiconductor switches  34   a ,  34   b  are jointly driven by the pulse phase modulator  36 , and are thus switched substantially simultaneously. The terminals of the transmitter coil  30  are furthermore connected to a bridge circuit  38  connected in parallel with the resonant circuit  28 . 
     The bridge circuit  38  is embodied as an H-bridge circuit having two bridge branches  40  each have two semiconductor switches  42   a ,  42   b ,  42   c ,  42   d . In this case, the semiconductor switches  42   a ,  42   b ,  42   c ,  42   d  are driven in a pulse-width-modulated manner. At one end in each case the bridge branches  40  are connected to a supply voltage by way of a potential terminal  44 . At the other end the bridge branches  40  are contacted with a ground potential by way of a second potential terminal  46 . Via the semiconductor switches  42   a ,  42   b ,  42   c ,  42   d  the respective coil end of the transmitter coil  30  is connectable either to the supply voltage or to the ground potential. By way of example, if the semiconductor switch  42   a  is closed (conducting) and the semiconductor switch  42   b  is opened (nonconducting), then the coil end coupled to the capacitor  32   a  is connected to the potential of the supply voltage. Correspondingly, in the event of the semiconductor switch  42   b  being opened and the semiconductor switch  42   a  being closed, the transmitter coil  30  is contacted with the ground potential. 
     The bridge circuit  38  is provided and set up for feeding loss energy (on account of parasitic losses or radiated power) into the resonant circuit  28  during operation of the transmitter  22 , in order to obtain a constant amplitude in the resonant circuit  28 —and thus a constant amplitude of the transmission signal  26 . 
     By means of the semiconductor switches  34   a ,  34   b , the resonant circuit  28  is started and halted or stopped. In this case, a switch-off time of the semiconductor switches  34   a ,  34   b  which is as accurate as possible is critical for the operation of the transmitter circuit  24 . The semiconductor switches  34   a ,  34   b  are suitably opened when the entire energy of the resonant circuit  28  is stored in the capacitors  32   a ,  32   b  and there is no residual energy in the transmitter coil  30 . This ensures that the transmitter circuit  24  is operated with as maximal an efficiency as possible since any residual energy in the transmitter coil  30  would be lost upon the opening of the semiconductor switches  34   a ,  34   b . Consequently, voltage spikes at the transmitter coil  30  (self-induction) are thus avoided, which voltage spikes could lead to undesired transmissions or even to the destruction of the transmitter circuit  24 . These effects cannot be completely avoided, however, on account of possible asymmetries in the transmitter circuit  24  or on account of a detuned resonant circuit  28 . The transmitter coil  30  is therefore short-circuited when the semiconductor switches  34   a ,  34   b  are open in order that the energy remaining in the transmitter coil  30  is short-circuited. The short-circuiting is effected either by means of a separate switch or by means of corresponding driving of the bridge circuit  38 , for example by means of an activation of the semiconductor switches  42   b  and  42   d  when the semiconductor switches  34   a  and  34   b  are open. 
     Preferably, the operating or switching states of the bridge circuit  38  and of the pulse phase modulator  36  are derived from a common clock generator or timer. This ensures synchronized switching of the semiconductor switches  34   a ,  34   b ,  42   a ,  42   b ,  42   c ,  42   d . Correct points in time of switching are thus ensured, such that the resonant circuit  28  can operate in resonance at a desired transmission frequency f 0  without the need to monitor an actuating voltage in the capacitors  32   a ,  32   b  or a coil current Is in the transmitter coil  30 . As a result, additional ammeters and/or voltmeters are obviated, thereby ensuring a simple, cost-effective and, in terms of structural space, compact construction of the transmitter circuit  24 . 
     The diagram in  FIG. 4  contains three sections  48 ,  50 ,  52  arranged vertically one above another. 
     The section  48  shows a schematic time-coil current diagram, wherein time t is plotted horizontally, i.e. along the abscissa axis (X-axis), and the coil current Is of the transmitter coil  30  is plotted along the vertical ordinate axis (Y-axis). 
     The section  50  shows the profile of a control signal of the pulse phase modulator  36  for the semiconductor switches  34   a ,  34   b , with the section  52  showing the control signals for the semiconductor switches  42   a ,  42   b ,  42   c ,  42   d . A high signal level of the control signals closes the respective semiconductor switch  34   a ,  34   b ,  42   a ,  42   b ,  42   c ,  42   d , i.e. switches it to the conducting state, a low signal level of the control signal corresponding to opening the respective semiconductor switch  34   a ,  34   b ,  42   a ,  42   b ,  42   c ,  42   d , i.e. switching it to the nonconducting state. In this case, the semiconductor switches  42   a  and  42   c  are opened and the semiconductor switches  42   b  and  42   d  are closed, while the semiconductor switches  34   a ,  34   b  are open in order that the transmitter coil  30  is short-circuited. 
     The section  48  schematically shows a modulation according to the invention of the transmission signal  26 . In this case, the transmission signal  26  is shown as a sinusoidal signal. The section  48  shows five transmitter patterns (TX sample)  54   a ,  54   b ,  54   c ,  54   d ,  54   e  for realizing different modulation phases. By way of example, in this case, provision is made of five sine waves per transmitter pattern  54   a ,  54   b ,  54   c ,  54   d ,  54   e , with one sine wave being omitted in the case of positive phase shifts (rotating in the counterclockwise direction), which results in four sine waves in these cases. 
     In the case of the transmitter pattern  54   a , a modulation phase of 0° is present, the transmission signal  26  having five sine waves. In the transmitter pattern  54   b , a phase shift of −90° is present, in which case the phase modulator  36  opens the semiconductor switches  34   a ,  34   b  in order to momentarily stop or pause the resonant circuit  28  until the desired phase angle of the sinusoidal signal is present. The subsequent transmitter pattern  54   c  likewise has a phase shift of −90°, but only four sine waves with a pause corresponding to a +90° phase shift are provided in order to realize again a phase angle or phase shift of 0° (relative to the transmitter pattern  54   a ) in the transmitter pattern  54   d . During the transmitter pattern  54   d , the resonant circuit  28  is paused in order to realize a phase shift of +135° for the transmitter pattern  54   e . Since the phase +135° is transmitted in the case of the transmitter pattern  54   e , it is necessary for the sinusoidal oscillation to be started before the actual symbol limit in order to attain a phase of 135°. In this case, the fourth transmitter pattern  54   d  has only four sine waves (sine waves with phase of 0°). 
     The pulse phase modulator  36  has the disadvantage that there is no continuous TX signal since it has to be halted or stopped in the case of phase shifts. This causes signal emissions outside a desired frequency band, too, which are typically higher than in the case of continuous operation. In order to reduce these undesired emissions, switching dithering is provided. 
     With reference to  FIG. 5 , a more detailed explanation is given below of switching dithering of the pulse phase modulator  36 , wherein the drive signals for the semiconductor switches  34   a ,  34   b  are provided with dithering noise. 
     The diagram in  FIG. 5  comprises two sections  56 ,  58  arranged vertically one above the other. Sections  56 ,  58  each show a schematic time-coil current diagram, wherein time t is plotted horizontally, i.e. along the abscissa axis (X-axis), and the coil current Is of the transmitter coil  30  is plotted along the vertical ordinate axis (Y-axis). In this case, the time axis is subdivided into phase angles. 
     In this case, the semiconductor switches  34   a ,  34   b  are opened when the capacitors  32   a ,  32   b  attain a positive or negative voltage maximum, which corresponds to two switching processes during a sine wave at 0° and 180°. In the differential construction of the transmitter circuit  24 , the charge in this case is exchanged between the two capacitors. Here and hereinafter, a switching position at 0° should be understood to mean in particular the state when the capacitor  32   a  has maximum voltage (the complete charge) and the other capacitor  32   b  has no voltage. The other point in time of switching at 180° is correspondingly the opposite, when the capacitor  32   b  has maximum voltage and the capacitor  32   a  has no voltage. 
     Overall, therefore, three different positions are possible for the restart of the resonant circuit  28  for a predefined TX phase. 
     First, a start in the case of a desired TX phase with a positive half-cycle. This point in time is provided with the reference sign  60  in  FIG. 4 . In the section  56 , the point in time  60  corresponds to a TX phase of 135°, with a TX phase of 337.5° being chosen in the section  58 . The resulting signal profile is provided with the reference sign  62  in the figures. 
     As a second start position, a start with one half-cycle earlier than at the point in time  60 , i.e. a start with a negative half-cycle but 180° earlier phase angle, is possible. This variant is shown in the section  56 , in the case of which variant the resonant circuit  28  is started at an earlier point in time  62 , thus resulting in a signal profile  66 . As of the point in time  60  the signal profiles  62  and  66  have the same temporal profile. 
     As a third start position, a start with one half-cycle later than at the point in time  60 , i.e. a start with a negative half-cycle but 180° later phase angle, is possible. This variant is shown in the section  58 , in the case of which variant the resonant circuit  28  is started at a later point in time  68 , thus resulting in a signal profile  70 . As of the point in time  68  the signal profiles  62  and  66  have the same temporal profile. The switch-off point in time is suitably chosen according to the planned start position. For a start with a positive half-cycle 0° it is necessary for the resonant circuit  28  to be stopped previously such that all charge is stored in one of the capacitors  32   a ,  32   b , for example in the capacitor  32   a . For a start with a negative half-cycle 180° it is accordingly necessary for the resonant circuit  28  to be stopped such that the charge is stored in the respective other capacitor, i.e. the capacitor  32   b , for example, in order that the coil current Is can start with a negative half-cycle. 
     Preference is given to changing over randomly between these three start or switching positions, as a result of which undesired transmissions are significantly reduced and the length of the transmitted phases is equalized. This dithering or dither noise is able to be realized by means of the following pseudocode, for example: 
     i) if (rand&gt;0.5):
 
ii) if (txPhase&gt;180°)
 
iii) startPosition=Dither_neg_late;
 
iv) else
 
v) startPosition=Dither_neg_early;
 
vi) end
 
vii) else
 
viii) startPosition=Default_pos;
 
ix) end
 
     In this case, rand is a random number between zero (0) and one (1), wherein txPhase is the transceiver phase in a range of between 0° and 360°. The switching position is designated by startPosition, where the start position without a delay with a positive half-cycle is designated as Default_pos, and the start position with 180° earlier negative half-cycle is designated as Dither_neg_early, and the start position with 180° later negative half-cycle is designated as Dither_neg_late. 
     In addition or as an alternative to switching dithering, phase polarity dithering is also conceivable, for example. In this case, the switch-off periods are shortened by the resonant circuit  28  being restarted with a phase offset of 0° or 180°. The resonant circuit is preferably started with 0° or 180°, irrespective of whether it was stopped previously at 0° or 180°. The polarity dithering thus differs from the exemplary embodiment described above in particular to the effect that the exemplary embodiment described above has to be switched off at 0° in order also to be able to be switched on again at 0°. In the case of this refinement by means of an additional H-bridge, it is possible to switch off at 180° but to switch on again at 0° (and vice versa) which shortens the switch-off periods. Furthermore, spectrum emissions are reduced and the transmitted energy is increased. In this case, the additional H-bridge connected between the capacitors  32   a ,  32   b  and the transmitter coil  30  enables a rotation or inversion of the polarity of the coil current Is, which enables a restart of the resonant circuit  28  with opposite polarity, i.e. a phase offset of 180°. As a result, the maximum possible gap or pause of the stopped resonant circuit  28  decreases from one full sine wave down to half a sine wave. 
     A further aspect of the pulse phase modulator  36  presented is the possibility of realizing 1-bit amplitude control. In this case, the resonant circuit  28  is stopped for example not just until the desired TX phase is attained, but also for a longer period of time that enables a “transmission” amplitude of the transmission signal  26  of zero. The transmission spectrum can be improved further as a result. Furthermore, the performance of the entire communication system of the hearing device  2  is also improved, e.g. with the use of PSK transmitters (PSK: Phase Shift Keying). 
     Reducing interference emissions of the transmission signal  26  by means of a superposition of a plurality of symbol rates is explained in greater detail below with reference to  FIGS. 6 to 8 . 
       FIGS. 6 to 8  each show a frequency-amplitude diagram for a Fourier-transformed transmission signal  26 . In this case, a frequency fin megahertz (MHz), is plotted horizontally, i.e. along the abscissa axis (X-axis), and a normalized amplitude A of a Fast Fourier Transform (FFT) in decibels (dB) is plotted along the vertical ordinate axis (Y-axis). 
     During operation, the pulse phase modulator  36  generates relatively high first-order sidelobes  72  at f 0 ±1.5 fs, wherein f 0  is the carrier frequency or transmission frequency and fs is the symbol rate of the modulation. If the symbol rate fs is chosen to be very high in order to increase the data rate of the radio link, it can happen that the sidelobes  72  no longer lie within permissible legal bandwidth limits, and must therefore be damped to a signal level that is predefined by an emission mask  74 . 
       FIG. 6  shows one example of the contravention of the emission mask  74  according to the European standard ETSI EN 300 330 V2.1.1. The allowed modulation in this case must lie within the bandwidth of ±7.5%×f 0 , and the interference emission level to the left and right thereof in this case must be damped by approximately 15 dB. 
     In order to reduce the interference emissions, provision is made for the symbol rate fs of the transmission signal  26  to be switched over between at least two symbol rate values fs 1 , fs 2 . In other words, the symbol rate fs of the transmission signal  26  is not constant, but rather changes back and forth between two or more symbol rates fs 1 , fs 2 . This effectively results in two transmission signals  26   a ,  26   b . In  FIG. 7 , the symbol rate fs 1  is slightly less than the symbol rate fs, with the symbol rate fs 2  being slightly greater than the symbol rate fs. Preferably, the symbol rates fs 1 , fs 2  differ from the transmission frequency f 0  by 20% or less in this case. In the exemplary embodiment, the transmission frequency is approximately 3.28 MHz (megahertz), with the symbol rate fs being approximately 281 kHz (kilohertz), the symbol rate fs 1  being approximately 234 kHz, and the symbol rate fs 2  being approximately 328 kHz. The symbol rate fs 1  is thus approximately 17% greater than fs, with the symbol rate fs 2  being correspondingly approximately 17% less than fs. 
     In the case where the symbol rates fs 1 , fs 2  are switched over frequently or rapidly, the emission spectra  26   a ,  26   b  are smeared to form a combined spectrum  26 ′ that no longer contravenes the prescribed emission mask  74 .  FIG. 8  shows the resulting spectrum or transmission signal  26 ′ resulting from the superposition of the different symbol rates fs 1 , fs 2  or transmission signals  26   a ,  26   b . The superposition of the different symbol rates fs 1 , fs 2  suppresses the sidelobes  72 . In this case, the switchover or changeover sequence between the symbol rates f 1 , f 2  is stored for the signal connection  6  or e2e communication both at the transmitter end and at the receiver end. 
     The invention is not restricted to the exemplary embodiments described above. Rather, other variants of the invention can also be derived therefrom by the person skilled in the art without departing from the subject matter of the invention. In particular, furthermore, all individual features described in association with the exemplary embodiments are also combinable with one another in a different way, without departing from the subject matter of the invention. 
     The following is a summary list of reference numerals and the corresponding structure used in the above description of the invention:
       2  Hearing device     4   a ,  4   b  Individual device     6  Signal connection     8  Device housing     10  Input transducer     12  Lines     14  Signal processing facility     16  Line     18  Receiver     20  Battery     22  Transmitter     24  Transmitter circuit     26 ,  26   a ,  26   b ,  26 ′ Transmission signal     28  Resonant circuit     30  Transmitter coil     32 ,  32   a ,  32   b  Capacitor     34 ,  34   a ,  34   b  Semiconductor switch     36  Pulse phase modulator     38  Bridge circuit     40  Bridge branch     42   a ,  42   b ,  42   c ,  42   d  Semiconductor switch     44 ,  46  Potential terminal     48 ,  50 ,  52  Section     54   a ,  54   b ,  54   c ,  54   d ,  54   e  Transmitter pattern     56 ,  58  Section     60  Point in time     62  Signal profile     64  Point in time     66  Signal profile     68  Point in time     70  Signal profile     72  Sidelobe     74  Emission mask   f 0  Transmission frequency   Is Coil current   t Time   f Frequency   fs, fs 1 , fs 2  Symbol rate