Patent Publication Number: US-6340956-B1

Title: Collapsible impulse radiating antenna

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of the filing of U.S. Provisional Patent Application Serial No. 60/165,084, entitled Collapsible Impulse Radiating Antennas, filed on Nov. 12, 1999, and the specification thereof is incorporated herein by reference. 
    
    
     GOVERNMENT RIGHTS 
     The U.S. Government has a paid-up license in this invention as provided for by the terms of SBIR Contract No. F29601-98-C-0004 awarded by the U.S. Air Force. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention (Technical Field) 
     The present invention relates to the field of impulse radiating antennas, specifically to wideband collapsible and portable impulse radiating antennas for ease of transport and deployment in the field. 
     2. Background Art 
     Note that the following discussion refers to a number of publications by author(s) and year of publication, and that due to recent publication dates certain publications are not to be considered as prior art vis-a-vis the present invention. Discussion of such publications herein is given for more complete background and is not to be construed as an admission that such publications are prior art for patentability determination purposes. 
     The present invention is a collapsible impulse radiating antenna (“CIRA”), which is a compact and lightweight implementation of the general class of antennas known as impulse radiating antennas (IRAs). IRAs are well suited for radiating an extremely broad band of signal frequencies at reasonable gain throughout the band. While the antenna gain is not optimal at any one frequency, it is sufficient for many applications over frequency ranges of around two decades (100:1 frequency ratio). Such devices also provide the ability to radiate an impulse-like electric field, when driven by a step-like voltage. Furthermore such devices are typically well matched to a 50-ohm impedance, so there is little power lost due to reflection from the antenna back into the source. Reflector IRAs generally consist of a parabolic reflector with a transverse electromagnetic (TEM) feed resulting in very broadband performance (2 decades) with a very narrow beam. 
     IRAs are useful in a wide variety of applications, including broadband communications and broadband radar. Broadband communications may include two distinct types of communication. First, broadband communications include conventional narrowband communications that are swept in frequency over large bandwidths. As an example, one may wish to listen to a very broad range of frequencies (or radio channels) without changing antennas. Second, broadband communications may include the radiation or reception of instantaneously broadband signals, which are often impulse-like in shape. This mode of communication is primarily digital, and is commonly implemented with pulse position modulation. In this form of modulation, a one or a zero is interpreted based on the time of arrival of an impulse relative to some time standard. 
     Broadband radar, like broadband communications, can encompass methods that require the use of either narrowband signals that are swept over a broad frequency band, or the use of signals that are instantaneously broadband or impulse-like. Broadband radar can have applications in the detection of mines or unexploded ordnance. It can also have application in the detection of cracks in road beds or in bridges. Furthermore, it can have applications in target identification, where the broad bandwidth is utilized to provide more information than what is normally generated by a narrowband radar system. Finally, broadband radar can be useful in Synthetic Aperture Radar (SAR), which can be used to map out ground features from the air. 
     An IRA enables a single antenna to perform multiple narrowband missions on a platform, such as a ship or satellite, with limited space available for antennas. While each of the missions may be intrinsically narrowband, the combined mission of the platform may require each of them to share a single broadband antenna. 
     Any of the IRA applications described above may, at times, require a portable version of the IRA to enable practical system development. This will occur if a system requires both high gain and portability. High gain forces one to use a large antenna, while portability suggests a small design. IRAs are generally fabricated from a solid reflector, which is clumsy to deploy and transport particularly when it reaches a certain size. 
     Several issued patents address the need for portable antennas and describe various collapsible configurations, some of which allow for stowing and deploying a paraboloidal reflector. None of these patents include the features of a broadband feed enabling a broad bandwidth for the antenna, collapsibility, and portability. U.S. Pat. No. 3,707,720 entitled, “Erectable Space Antenna” to Staehlin et al. describes a collapsible antenna for use in space. The antenna described is not applicable for a large bandwidth or for ultra-wide band use; the reflector is flat and cannot achieve a paraboloidal shape thereby compromising the available gain. 
     U.S. Pat. No. 4,642,652 to Herbig et al., entitled, “Unfoldable Antenna Reflector” discloses a collapsible antenna wherein bracing wires placed behind the antenna are used to provide the tension force to maintain the antenna&#39;s shape. U.S. Pat. No. 5,963,182 to Bassily entitled, “Edge-Supported Umbrella Reflector with Low Stowage Profile” discloses an umbrella-type antenna for use on a spacecraft where the ribs of the antenna are fixed in a parabolic shape using a rigid truss structure. U.S. Pat. No. 5,635,946 to Francis entitled, “Stowable, Deployable, Retractable Antenna” discloses a retractable and deployable antenna wherein cables are used to deploy as well as support the reflector. U.S. Pat. No. 4,899,167 to Westphal entitled, “Collapsible Antenna” discloses a collapsible antenna where rigid saw-tooth shaped segments collapse into one another to collapse the reflector. U.S. Pat. No. 3,618,111 to Vaughan entitled, “Expandable Truss Paraboloidal Antenna” discloses a collapsible antenna made up of a plurality of interconnecting hinged solid triangular supports making up a truss antenna structure. U.S. Pat. No. 3,982,248 to Archer entitled, “Compliant Mesh Structure for Collapsible Reflector” discloses a collapsible antenna made of a wire mesh structure with spring-loaded wires that expand to a certain shape when deployed. The elasticity of the mesh allows the material to take shape when deployed. U.S. Pat. No. 4,295,143 to Winegard et al. entitled, “Low Wind Load Modified Parabolic Antenna” discloses a collapsible reflector boom having two parabolic reflectors mounted thereon. Solid reflector elements make up the two symmetrical parabolic reflectors. 
     None of the antennas described in the above patents provide a lightweight, portable, ultra-wideband collapsible antenna. The present invention for a collapsible impulse radiating antenna overcomes the deficiencies in the prior art patents by providing a high gain, ultra-wideband antenna that comprises a reflector made of a conductive mesh fabric that is lightweight and collapsible in an easy umbrella-like fashion. The present invention enables all of the applications discussed above and many others, because it is more portable and lightweight than conventional IRAs. In the preferred embodiment, the present invention for a collapsible IRA (“CIRA”) weighs only five pounds and is about the size of a typical umbrella, making it easily transportable by an individual, and easily deployable in the field. 
     In a second embodiment, the CIRA includes expandable seams between adjacent panels of the reflector, enabling the reflector surface curvature to be adjusted from a more focused to a less focused mode. The flexibility of this embodiment provides a collapsible multifunction IRA (“CMIRA”). 
     SUMMARY OF THE INVENTION (DISCLOSURE OF THE INVENTION) 
     The present invention is a broadband collapsible impulse radiating antenna having a reflector and feed arms made from a flexible conductive material. The antenna is operational over a broad bandwidth, in a range from below 50 MHz to above 8 GHz. When driven by a step function, the antenna can radiate an impulse on boresight having a full-width-half-maximum of less than one-fifth the time required for light to travel a distance of one reflector diameter in free space. An umbrella-like support mechanism is used to collapse and deploy the reflector. The umbrella-like mechanism consists of a plurality of support ribs, a center support rod, center push rods, feed arm support rods, and a push sleeve. The support ribs are attached to the reflector and are pivotally connected to a central hub and pivot radially inward and outward upon collapsing and deploying the antenna. A push sleeve slides along the center support rod causing the radial center push rods, that pivot at the push sleeve as well as at the reflector, to provide a radial force to the reflector and thereby deploy and collapse the antenna. A center can maintains the center support rod in a fixed position and contains an RF splitter that splits the input signal into two feed cables of equal length leading to the feed point. Expandable seams are optionally provided in the reflector and feed arms so that the surface curvature of the reflector can be adjusted. The antenna is lightweight, weighing less than three pounds per foot of reflector diameter. 
     A primary object of the present invention is to provide a collapsible broadband IRA antenna that is easily deployed in the field. 
     A primary advantage of the present invention is that it is compact, lightweight, and can be easily transported and deployed in the field by a single individual. 
     Other objects, advantages and novel features, and further scope of applicability of the present invention will be set forth in part in the detailed description to follow, taken in conjunction with the accompanying drawings, and in part will become apparent to those skilled in the art upon examination of the following, or may be learned by practice of the invention. The objects and advantages of the invention may be realized and attained by means of the instrumentalities and combinations particularly pointed out in the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings, which are incorporated into and form a part of the specification, illustrate several embodiments of the present invention and, together with the description, serve to explain the principles of the invention. The drawings are only for the purpose of illustrating a preferred embodiment of the invention and are not to be construed as limiting the invention. In the drawings: 
     FIG. 1 is a frontal perspective view of the preferred embodiment of the present invention showing the CIRA in the deployed (a) and collapsed (b) positions; 
     FIG. 2 is a rear view of FIG. 1 showing the backside of the reflector in the deployed (a) and collapsed (b) positions; 
     FIG. 3 is a front view of the CIRA of FIG. 1 in the deployed position; 
     FIG. 4 is a cross-section of the CIRA taken along line  50 — 50 , across two diametrically disposed feed arms, of FIG. 3; 
     FIG. 5 is a cross-section of the CIRA taken from  50  to the vertex of the reflector of FIG. 3, along one feed arm, with the reflector removed from view; 
     FIG. 6 is a cross-section of the CIRA taken along line  60 — 60  in FIG. 3, the direction of the dominant polarization of the CIRA, a full diameter of the reflector of FIG.  3  through the midpoint of two diametrically disposed panels of the reflector; 
     FIG. 7 is a side view of the CIRA of FIG. 1 in the deployed position; 
     FIG. 8 is a close-up perspective exploded view of the feed point area of the present invention showing the feed point area cover and feed point support flange, the center support rod, and four feed arms in the deployed position; 
     FIG. 9 is close-up perspective cutaway view of the center can, and backside of the reflector of the CIRA of FIG. 1 in the deployed position; 
     FIG. 10 a  is a cross-sectional schematic diagram of the splitter used in accordance with the present invention showing the center feed cable running through the center support rod and the radial feed cable running along one of the feed arms; the diagram illustrating the principle that both cables are of equal lengths from the RF splitter and connected in parallel from the center can to the feed point of the antenna; 
     FIG. 10 b  is a schematic top view of the center and radial feed cables of FIG. 10 a  leading to the feed point of the antenna; 
     FIG. 10 c  is a close-up view of the feed point of FIG. 10 b , showing the feed cable connections at the feed point; 
     FIG. 11 is a frontal perspective view of the CMIRA embodiment of the present invention shown in the focused (a), defocused (b), and collapsed (c) positions; 
     FIG. 12 is a cross-section of FIG. 11 in the defocused deployed position through two diametrically disposed feed arms; 
     FIG. 13 is a close-up perspective view of an expandable seam used to adjust the surface curvature and thereby adjust the focus of the reflector of the CMIRA of FIG. 17; 
     FIG. 14 is a close-up perspective view of an expandable seam on a feed arm used when adjusting the length of a feed arm of the CMIRA as needed when adjusting the surface curvature of the reflector of the CMIRA; 
     FIG. 15 is a schematic diagram of the antenna data acquisition system used for measuring the characteristics of the antenna of the present invention; 
     FIG. 16 a  is the time domain reflectometry plot of the FRI-TEM-02-100, a 100Ω half TEM horn mounted against a truncated ground plane used in the data acquisition system of FIG. 15 when measuring the characteristics of the ultra-lightweight CIRA configuration of the present invention; 
     FIG. 16 b  is a plot of the normalized impulse response in the time domain of the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 16 c  is a plot of the normalized impulse response in the frequency domain of the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 16 d  is a plot of the IEEE standard gain in the frequency domain of the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 17 a  is the time domain reflectometry plot of the FRI-TEM-01-50, a 50Ω half TEM horn mounted against a truncated ground plane used in the data acquisition system of FIG. 15 when measuring the ultra-lightweight CIRA configuration of the present invention; 
     FIG. 17 b  is a plot of the normalized impulse response of the of the plot of FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 17 c  is a plot of the normalized impulse response in the frequency domain of the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 17 d  is a plot of the IEEE standard gain in the frequency domain of the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 18 a  is the time domain reflectometry plot of the ultra-lightweight configuration of the CIRA of the present invention showing the connector, splitter, feed cable, feed point, and resistors; 
     FIG. 18 b  is a plot of the raw impulse response data on boresight with the ground bounce removed and zero-padded of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 18 c  is a plot of the raw impulse response data on boresight with the ground bounce removed and zero-padded of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 18 d  is a plot of the expanded normalized impulse response in the time domain of FIG. 18 b  wherein the FWHM=73 picoseconds; 
     FIG. 18 e  is a plot of the expanded normalized impulse response in the time domain of FIG. 18 c  wherein the FWHM=68 picoseconds; 
     FIG. 18 f  is a plot of the normalized impulse response in the frequency domain of FIG. 18 b;    
     FIG. 18 g  is a plot of the normalized impulse response in the frequency domain of FIG. 18 c;    
     FIG. 18 h  is a plot of the IEEE standard gain in the frequency domain of the ultra-lightweight configuration of the CIRA of the present invention on boresight acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 18 i  is a plot of the IEEE standard gain in the frequency domain of the ultra-lightweight configuration of the CIRA of the present invention on boresight acquired with the data acquisition system of FIG. 15 with the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 18 j  is a plot of the conventional impulse response of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 18 k  is a plot of the conventional impulse response of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 18 l  is a plot of the integrated impulse response of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 18 m  is a plot of the integrated impulse response of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-01-50 sensor of FIG. 17 a;    
     FIG. 19 a  is a plot of the raw data cross polarization response of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 19 b  is a plot of the IEEE gain on boresight of the cross polarization response of FIG. 19 a;    
     FIGS. 20 a  through  20   h  show the IEEE gain plotted as a function of the angle off-boresight in the H-plane for the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a , wherein plots are provided at 98 MHz, 195 MHz, 391 MHz, 586 MHz, 781 MHz, 977 MHz, 2,002 MHz, and 4,004 MHz; 
     FIGS. 21 a  through  21   h  show the IEEE gain plotted as a function of the angle off-boresight in the E-plane for the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a , wherein plots are provided at 98 MHz, 195 MHz, 391 MHz, 586 MHz, 781 MHz, 977 MHz, 2,002 MHz, and 4,004 MHz; 
     FIG. 22 is the antenna pattern based on peak raw voltage measurements in the H-plane as a function of the angle off-boresight of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 23 is a plot of raw voltages at several angles in the H-plane as a function of time of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 24 is the antenna pattern based on peak raw voltage measurements in the E-plane as a function of the angle below boresight of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 25 is a plot of raw voltages at several angles in the E-plane as a function of time of the ultra-lightweight configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 16 a;    
     FIG. 26 is a perspective view of the FRI-TEM-02-100, a 100Ω half TEM horn mounted against a truncated ground plane used in the antenna measurement configuration of FIG. 15; 
     FIG. 27 is a perspective view of the CIRA of the present invention in the collapsed position (c) shown alongside a tripod (b) for mounting the CIRA upon and a clamp (a) for clamping the CIRA to other objects in the field; 
     FIG. 28 is a close-up perspective view of the CIRA clamped to a fence post in the field with the clamp of FIG. 27 a;    
     FIG. 29 is a side view of the CIRA mounted on the tripod of FIG. 27 b;    
     FIG. 30 a  is the time domain reflectometry plot of an early model of the FRI-TEM-02-100, a 100Ω half TEM horn mounted against a truncated ground plane used in the antenna measurement configuration of FIG. 15 when measuring the characteristics of a 20-panel CIRA configuration of the present invention; 
     FIG. 30 b  is a plot of the normalized impulse response in the time domain of the FRI-TEM-02-100 sensor of FIG. 30 a , wherein the FWHM=52 picoseconds; 
     FIG. 30 c  is a plot of the normalized impulse response in the frequency domain of the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 31 a  is a plot of the normalized impulse response in the time domain of a 20-panel configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a , wherein the FWHM=105 picoseconds; 
     FIG. 31 b  is a plot of the normalized impulse response on boresight in the frequency domain of the 20-panel configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 31 c  is a plot of the IEEE gain of the 20-panel configuration of the CIRA of the present invention on boresight acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 32 is the antenna pattern based on peak raw voltage measurements in the H-plane as a function of the angle off-boresight of the 20-panel configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 33 is the antenna pattern based on peak raw voltage measurements in the E-plane as a function of the angle below boresight of the 20-panel configuration of the CIRA of the present invention acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 34 a  is a plot of the normalized impulse response in the time domain of a 20-panel CMIRA configuration in the focused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 34 b  is a plot of the normalized impulse response on boresight in the frequency domain of the 20-panel CMIRA configuration in the focused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 34 c  is a plot of the IEEE gain on boresight of the 20-panel CMIRA configuration in the focused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 35 is the antenna pattern based on peak raw voltage measurements in the H-plane as a function of the angle off-boresight of the 20-panel CMIRA configuration in the focused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 36 is the antenna pattern based on peak raw voltage measurements in the E-plane as a function of the angle below boresight of the 20-panel CMIRA configuration in the focused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 37 a  is a plot of the normalized impulse response in the time domain of the 20-panel CMIRA configuration in the defocused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 37 b  is a plot of the normalized impulse response on boresight in the frequency domain of the 20-panel CMIRA configuration in the defocused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 37 c  is a plot of the IEEE gain on boresight of the 20-panel CMIRA configuration in the defocused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 38 is the antenna pattern based on peak raw voltage measurements in the H-plane as a function of the angle off-boresight of the 20-panel CMIRA configuration in the defocused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a;    
     FIG. 39 is the antenna pattern based on peak raw voltage measurements in the E-plane as a function of the angle below boresight of the 20-panel CMIRA configuration in the defocused mode acquired with the data acquisition system of FIG. 15 with the FRI-TEM-02-100 sensor of FIG. 30 a ; and 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS (BEST MODES FOR CARRYING OUT THE INVENTION) 
     The present invention for a collapsible IRA provides both broadband performance along with portability. The antenna is collapsed and deployed in an umbrella-like fashion, having a reflector and feed arms sewn from flexible conductive and resistive fabric. There are two basic embodiments of the invention; the first embodiment is referred to herein as the Collapsible IRA, or “CIRA”. The second embodiment has expansion seams in the reflector to allow the surface curvature to be adjustable and is referred to herein as the Collapsible Multifunction IRA, or “CMIRA”. The CMIRA is a multifunction antenna due to the adjustable surface curvature of the reflector providing more and less focused modes of operation and an adjustable beamwidth as needed for the particular task. 
     The preferred embodiment of the present invention for a collapsible IRA is shown in FIGS. 1-9. First, the basic antenna elements will be described. FIG. 1 is a frontal perspective view showing the CIRA  10  in the deployed (FIG. 1 a ) and collapsed (FIG. 1 b ) positions. Turning to FIG. 1 a  showing CIRA  10  in the deployed position, it can be seen that reflector  36  is made up of a plurality of truncated, roughly-triangular shaped panels  42  (more easily seen in FIGS. 2 and 3) which are made from a flexible conductive material. Preferably, reflector  36  is made of between 12 and 20 triangular panels  42  connected together with the smaller ends of each triangular panel connecting to a common central circular panel  44  (see also FIG. 2) to form a paraboloid. Twelve triangular panels  42  are shown in the figures (See FIGS. 2 and 3.); however, the number, shape and arrangement of the panels can of course vary so long as the reflector surface curvature is not compromised beyond that necessary to achieve the desired antenna characteristics. As an example, a reflector that is a 1.22 meter (48 inch) diameter parabolic dish with a focal length of 0.488 meters (F/D=0.4) with an upper frequency of interest of 2 GHz, should in theory have a reflector surface variation of only about 7.6 mm from a true paraboloid, which is only about 5% of the wavelength at 2 GHz. The depth of this example reflector would ideally be 190 mm. 
     Center support rod  22  extends a distance away from the vertex of and along the axis of symmetry of reflector  36  and provides feed point  54  at the focal point of the paraboloidal reflector  36  as well as support for reflector  36  when in the deployed position. Focal length-to-diameter ratios (F/D) for IRAs are commonly between 0.25 and 0.5, inclusive. However, for the collapsible IRA of the present invention, a focal length-to-diameter ratio that is too long creates an antenna too large to satisfy the compact transportable nature desired. A focal length-to-diameter ratio that is too short creates an antenna that is difficult to deploy given the sharpness of the acute angle at which the push rods, which are described below, pivot to deploy the antenna. Consequently, a focal length of approximately 0.4 has been found to be a good compromise to achieve the desired characteristics of compactness and ease of deployment for the CIRA. 
     Four feed arms  24 ,  24 ′,  24 ″ and  24 ′″ extend from feed point  54  outward to the perimeter of reflector  36 , each shown ±45 degrees from the dominant polarization angle,  60 — 60 , of the antenna. (See FIG. 3.) However, IRAs with feed arms ±30 degrees from the dominant polarization, demonstrate improved cross-polarization (crosspol) purity which is the tendency of an antenna to radiate only the dominant polarization, without radiating the cross polarization. In many applications, such as in radar, it is disadvantageous to radiate crosspol, and therefore, the CIRA feed arms  24  can instead be positioned ±30 degrees from the dominant polarization angle. 
     While four feed arms  24  are shown, the present invention is not limited to this number of feed arms, as will be apparent to those skilled in the art of IRA design. Additional feed arms would require that each feed arm be narrower in order to maintain the same feed impedance. Feed arms  24  are each narrow at the feed point area, widen toward their midpoints, and then taper again at the ends which connect to reflector  36  in a diamond shape as can be seen in FIGS. 1,  4 ,  5 ,  6 , and  7 . Ideally the feed arms of an IRA extend from the feed point in a perfect triangular shape in order to have minimal reflection back into the source from the ends of the feed arms; however, this design is not always entirely practicable for the CIRA. Each of feed arms  24  can comprise a plurality of solid conductors, such as a plurality of wires extending from feed point  54  to reflector  36 , in an approximately parallel manner, being close together at feed point  54  and further apart as they get closer to reflector  36 , so that the wires together form a triangular shape. Feed arms  24  are arranged in diametrically disposed pairs, more easily seen when CIRA  10  is viewed from the front as shown in FIG.  3 . Each pair of feed arms  24  are oriented in a plane that includes the line formed by center support rod  22 . This is best seen in FIG. 4 which is a cross section of CIRA  10  taken along section-line  50 — 50  of FIG. 3, across two diametrically disposed feed arms. Each feed arm includes a resistive load  34 . The size and shape of resistive load  34  can vary but is generally located near reflector  36  in order to maintain a conductive triangular shaped feed arm as far as possible from the feed point, to maintain the fastest possible impulse response. 
     The antenna uses an umbrella-like mechanism to support reflector  36  and for deployment and collapsing reflector  36 . This mechanism includes support ribs  52 , described below, center support rod  22 , center push rods  28 , feed arm support rods  26 , and push sleeve  32 . In order to deploy CIRA  10  from the collapsed position shown in FIG. 1 b  to the deployed position shown in FIG. 1 a , the user grasps push sleeve  32  and slides it away from feed point  54  and along center support rod  22  toward circular panel  44 , thereby forcing each of a plurality of center push rods  28 , which are pivotally connected to center support rod  22 , to pivot at push rod pivot points  16  at the center of push sleeve  32  away from center support rod  22 . (See also FIGS. 4 and 5.) This action causes reflector  36  to expand from the collapsed position outward to the paraboloidal deployed position in an umbrella-like manner due to the far ends of center push rods  28  each being pivotally connected to reflector  36  at pivot points  31  in a concentric ring around circular panel  44  of reflector  36 . Additionally, feed arm support rods  26 , one for each feed arm  24 , are pivotally connected at points  30  to selected center push rods  28  at their reflector pivot points  31 , pivot points  30  and  31  sharing the same pivot. Feed arm support rods  26  are connected to feed arms  24  at the opposite end from reflector  36  by sliding into sleeves  33  located transversely to the length of and upon each of feed arms  24 , each sleeve being closed at the far end from where feed arm support rod  26  enters into sleeve  33 . The force of center push rods  28  moving radially outward upon deployment of CIRA  10  also forces feed arm support rods  26  to pivot at  30  radially outward away from center support rod  22 , and toward reflector  36  and thereby thrust feed arms  24  away from the collapsed position, where they are folded in toward center support rod  22 , to the deployed position, where they are fully extended between feed point  54  and the edges of reflector  36  in the deployed position. 
     The preferred catch mechanism for maintaining CIRA  10  in the deployed position operates by twisting push sleeve  32  a small amount upon reaching the desired position along center support rod  22 . After releasing pressure, push sleeve  32  then locks into a detent, thus maintaining pressure against push rods  28 . Another, less preferred catch mechanism is a nylon nut that is attached to push sleeve  32  and engages threads on center support rod  22  at the point at which reflector  36  is fully deployed, thereby providing the required force to fixedly hold push sleeve  32  and reflector  36  in the deployed position. Of course, other types of catch mechanisms can be used to fixedly hold push sleeve  32  in the deployed position as will be apparent to those skilled in the art. Optionally, center push sleeve  32  can be controlled by automatic mechanical means, such as a servo motor, allowing automatic deployment of CIRA  10  by electrical control. 
     In order to collapse CIRA  10 , push sleeve  32  is disengaged from its fixed position and slid in the reverse direction from deployment along center support rod  22  causing center support rods  28 , feed arm support rods  26 , and support ribs  52  (described next) to pivot in the opposite directions than during deployment. This action causes reflector  36  and feed arms  24  to collapse into a compact position as shown in FIGS. 1 b  and  2   b.    
     Turning to FIG. 2 for a rear view showing the backside of reflector  36  of CIRA  10  in the deployed position (FIG. 2 a ) and collapsed position (FIG. 2 b ), twelve triangular panels  42  can be seen connected to one another and to central circular panel  44 . Reflector  36  is supported in the deployed position by a semi-rigid frame made of a plurality of support ribs  52 , more easily seen in FIG. 5 which is a cross-section of CIRA  10  taken from  50  on FIG. 3 to the vertex of reflector  36  along one feed arm, with reflector  36  removed from view. Support ribs  52  run through mating sleeves  38  that are connected to the backside of reflector  36 . Support ribs  52  pivotally connect at points  56  near the vertex of reflector  36  to the end of center can  12  nearest reflector  36  via hub  66 . (See FIG. 9.) Pivot points  56  allow support ribs  52  to pivot in toward center support rod  22  upon CIRA  10  being collapsed and allow support ribs  52  to pivot radially outward away from center support rod  22  upon deployment of CIRA  10 . Center support rod  22  runs through the central axis of center can  12  and is fixedly held in can  12  via frame  48 . See FIG. 9 for a close-up perspective view of center can  12  (cut-away) and backside of reflector  36  showing support ribs  52  as they pivotally connect at  56  to the reflector end of can  12  as well as for a view of center support rod  22  centered within can  12 . 
     Support ribs  52 , as well as center push rods  28  and feed arm support rods  26  are made of a sufficiently rigid material to support reflector  36  in the deployed position. However, support ribs  52  are preferably flexible enough to allow deployment without excessive force required. In the preferred embodiment the CIRA can be deployed by a single person. A fiberglass reinforced material may be used for support ribs  52 . Support ribs  52  can be either conducting or nonconducting as they are located on the backside of reflector  36 . Push rods  28  and feed arm support rods  26  are nonconductive and have a low dielectric constant as close to that of free space as possible (for instance, approximately 2.5 or less). Push rods  28  need to be relatively strong to deploy and collapse reflector  36 , however support rods  26  do not require much strength as they are only supporting feed arms  24 . 
     FIG. 3 is a front view of CIRA  10  in the deployed position demonstrating the plurality of symmetrical center push rods  28  used to deploy reflector  36  as well as push rod pivot points  31  upon the front side of reflector  36 . FIG. 4 is a cross-sectional view of FIG. 3 taken along two diametrically disposed feed arms ( 50 — 50 ) showing center support rod  22  which runs through center can  12  and provides support for push sleeve  32  to slide upon in order to deploy reflector  36 . Center push rods  28  can be seen to pivot at pivot points  16  as push sleeve  32  slides along center support rod  22  during the deployment and collapsing actions. Turning to FIG. 5, center support rod  22  is held in place within can  12  by frame  48  (see also FIG.  9 ), which is affixed to can  12 . Each of support ribs  52  also provide a fixed connection point  58  for each of pivot points  31  through a small hole in reflector  36 . 
     FIG. 6 provides a cross-sectional view of FIG. 3 taken along  60 — 60 , the midpoints of two diametrically disposed triangular panels of CIRA  10 . It can be more easily seen in FIG. 6 that feed arms  24  connect to reflector  36  along radial lines, one of which is shown at  62 . FIG. 7 provides a side view of CIRA  10  revealing the backside of reflector  36  where mating sleeves  38  for the support ribs can be seen. A plurality of loops  40  are also provided (see also FIG. 2 a ) on the backside of reflector  36  for the attachment of ropes or cords making it possible to raise CIRA  10  into a tree or other structure and aim reflector  36  in a selected direction with the ropes. 
     Attention is now turned to FIG. 8 which is a close-up exploded view of feed point  54 . Feed point support flange  20  is comprised of four radial arms, 90 degrees apart, that are affixed to each of the four feed arms  24 ,  24 ′,  24 ″, and  24 ′″. Support flange  20  is preferably made of a material that is strong, has a low dielectric constant (such as approximately 2.8 ), is machinable, and has as small as possible an effect on the time domain reflectometry (TDR) of the antenna, such as a dielectric Ultra High Molecular Weight (UHMW) Polyethylene. Support flange  20  supports conductive tips  68  that are preferably made of copper, which are in turn attached to each of feed arms  24 , usually with an adhesive. Conductive tips  68  on feed arms  24  provide strength to feed arms  24  at the feed point and provide a means for electrically connecting the feed cables to the feed arms, such as by soldering. The feed cables will be described below. Feed point cover  18  fits over the apex of feed point  54  to provide mechanical support and to protect the electrical connections located at feed point  54 . Feed point cover  18  is also preferably made of a material that is strong, has a low dielectric constant (such as approximately 2.8), is machinable, and has as small as possible an effect on the TDR of the antenna. It can be seen in FIG. 8 that feed point cover  18  connects to support flange  20  by means of a series of mating screws and holes. Support flange  20  is affixed to feed arms  24  by screws and mating holes as well. These screws are nonconductive, and can be made from nylon. Other nonconductive attachment means for these feed point components will be apparent to those skilled in the art. 
     FIG. 9 provides a close-up view of center can  12 , cutaway, and the backside of reflector  36 . Reflector support ribs  52  can be seen within mating rib sleeves  38  on the backside of reflector  36  and pivotally connected at points  56  to the end of can  12  nearest reflector  36  via hub  66 . Center can  12  is shown cutaway to reveal center support rod  22  running longitudinally through the center of can  12  and supported therein by cylindrical frame  48 . Input port  70  is located on the outer surface of can  12  and is split within can  12  at  72  into two feed cables as will be described below. Tripod mount  14  is also shown connected to can  12  via knurled knob  64  which screws through mating holes in clamp  14  and the bottom of can  12  (not shown) and is held fixed by nut  74 . 
     FIG. 10 is a schematic diagram demonstrating the theory of the RF splitter used in accordance with the present invention. FIG. 10 a  is a schematic cross-sectional side view of reflector  36  showing feed cables  76  and  78  for feed point  54 . It is to be noted that splitter  86  is not shown in its usual location within center can  12  (not shown) at the vertex of and at the backside of reflector  36 , but is instead shown midway between feed cables  76  and  78  for purposes of demonstrating the equal lengths of the feed cables only. FIG. 10 b  is a front view of the reflector and FIG. 10 c  is a close-up front view of feed point  54 . The CIRA is fed by splitter  86  and two preferably 100-ohm feed cables  76  and  78  connected in a series/parallel manner, which have the effect of transforming a 50-ohm impedance at input port  70  into a 200-ohm impedance at feed point  54 . This is accomplished with minimal power loss due to reflection of signal from the antenna back into the source. Input port  70  can comprise a 50-ohm SMA connector, a 3-½ mm connector, a 7 mm connector, an N-type connector, or a variety of other high frequency connectors. The CIRA is normally fed at input port  70  with a standard 50-ohm cable for the best impedance match to standard equipment. Inside center can  12  (not shown), splitter  86  splits the input signal into two preferably 100-ohm cables,  76  and  78 , of equal length connected in parallel. Center cable  76  is fed up the center of hollow center support rod  22  toward feed point  54 . Radial cable  78  is attached to one of the support ribs  52  and then is fed along a feed arm  24  to reach feed point  54 . A ferrite bead is placed around radial cable  78  at the point where cable  78  crosses resistive load  34  (not shown) on feed arm  24  to prevent current on the exterior of cable  78  from shorting out resistive load  34 . Because the two cable lengths are the same, center cable  76  is longer than the physical distance from center can  12  to feed point  54  when center can  12  is located at the vertex of reflector  36 . Therefore the extra cable length of center cable  76  is taken up in windings within can  12  to absorb the extra cable length. 
     At feed point  54 , both cables  76  and  78  converge and are electrically connected to each other and to the four feed arms  24  as shown in FIG. 10 c , so that feed arms  24  and  24 ′ are connected to each other, and feed arms  24 ″ and  24 ′″ are connected to each other in such a manner as to provide the positive and negative terminals of the antenna to produce the electric fields. Cables  76  and  78  are connected at feed point  54  in a serial manner, so that their combined impedance is 200 ohms, which is a matched impedance to the antenna at feed point  54 . The net result is that a  50  ohm input impedance is transformed to a 200 ohm impedance at the feed point of the antenna. Cables  76  and  78  are connected in a manner minimizing effects on the TDR. 
     Attention is briefly drawn to FIGS. 27-29. FIG. 27 a  shows optional multi-purpose clamp  80  that is used when mounting CIRA  10  to objects in the field, such as a fence post as shown in FIG.  28 . FIG. 27 b  shows tripod  90 , preferably made of carbon fiber so that it is lightweight, that is also used to mount CIRA  10  as shown in FIG.  29 . Ball joint  84  is shown in use with clamp  80  and also with tripod  90  in FIGS. 28 and 29 to position and aim CIRA  10 . CIRA  10  is easily rotated to either horizontal or vertical polarization. Either of tripod  90  or clamp  80  connect to ball joint  84  which in turns connects to tripod mount  14  which is affixed to center can  12  as described above. Tripod mount  14  preferably provides a standard ⅜ inch-16 thread tripod connection. These devices are shown in FIG. 27 next to CIRA  10  in the collapsed position (FIG. 27 c ) to demonstrate the compact nature of the kit comprised of CIRA  10 , tripod  90 , ball joint  84 , and clamp  80 , which weighs less than twelve pounds altogether and can be easily transported into the field by a single individual in a backpack. Strap  46  is used to retain CIRA  10  in the collapsed position. In the preferred embodiment, the antenna can be set up in the field by one person and can be used with a variety of military and commercial off-the-shelf (COTS) transmitters and receivers. 
     Attention is returned to FIGS. 11-14 which provide further detail of the second embodiment of the present invention which is a multifunction version of CIRA  10  described above, and is referred to herein as CMIRA  100 . Reflector  102  of CMIRA  100  has an adjustable surface curvature and therefore has an adjustable beamwidth. It is to be understood that although two modes, focused and defocused, are discussed herein, CMIRA  100  can of course accommodate varying degrees of focus depending upon the degree of expansion of reflector  102  via expandable seams  106  discussed below. It is also to be understood that CMIRA  100  comprises the identical elements and operates in the identical fashion as CIRA  10  described above, but includes expandable seams in reflector  102  and feed arms  104 . All alternative and equivalent elements described with regard to CIRA  10  are equally applicable to CMIRA  100 . FIG. 11 a  shows CMIRA  100  in the deployed focused mode. When in the focused mode, reflector  102  of CMIRA  100  is a paraboloid and operates in the same fashion as CIRA  10  described above. FIG. 11 b  shows CMIRA  100  in the deployed defocused mode. The flatness of CMIRA  100  in FIG. 11 b  is exaggerated for purposes of demonstrating the difference between the focused and defocused modes of operation. FIG. 11 c  shows CMIRA  100  in the collapsed position. In the focused mode CMIRA  100  provides a narrower beamwidth and higher gain than in the defocused mode, thus making CMIRA  100  adaptable to more than one application. 
     FIG. 12 provides a side view of CMIRA  100  in the defocused mode, demonstrating that reflector  102  is more flattened. In order to be expanded to the defocused mode, reflector  102  is provided with a plurality of expandable seams  106  which can be seen in FIG. 13 with CMIRA  100  in the focused mode. Expandable seams  106  are located radially upon reflector  102  and comprise a triangular-shaped piece of conductive fabric, the narrow end of which is located radially closer to the vertex of reflector  102  and the wider end of which is located at the outer edge of reflector  102 . When in the focused mode, as shown in FIG. 11 a  and FIG. 13, each expandable seam  106  is held in a folded position by means of a conductive connector, such as mating Velcro connectors  108  and  110 , so that the integrity of the conductivity of reflector  102  is not compromised. Other types of conductive connectors can be used to maintain expandable seams  106  in the folded position, such as zippers with a conductive coating. Feed arms  104  are also provided with conductive expandable seams  112  as shown in FIG. 14 which are held in the folded position with mating conductive connectors  116  and  118  when reflector  102  is in the focused mode. Expandable seams  112  are rectangular-shaped and are located on feed arms  104  between the resistive loads and reflector  102 . Expandable seams  112  are preferably located as far as possible from the feed point in order to maintain the preferred triangular shape of the feed arms for as far as possible from the feed point. 
     To bring CMIRA  100  to the defocused mode as shown in FIG. 11 b  and FIG. 12, connectors  108  and  110 ,  116  and  118  are released thereby allowing the surface curvature of reflector  102  to adjust into a more flattened configuration, and allowing feed arms  104  to increase in length, due to the tension of the push rods against the support ribs of reflector  102 . Releasing connectors  108  and  110 ,  116  and  118  also provides continuous conductivity across reflector  102  and feed arms  104  through expandable seams  106  and  112 . 
     Push sleeve  120  is slid along center support rod  122  in the opposite direction, away from reflector  102 , to collapse CMIRA  100  into the position shown in FIG. 11 c  in the same manner described above with respect to the CIRA embodiment. 
     Center can  12 , frame  48 , and hub  66  (see FIG.  9 ), for both the CIRA and CMIRA embodiments are preferably strong and lightweight, and can comprise aluminum. The push sleeve is nonconductive, has a low dielectric constant, and is preferably made of a strong machinable material, such as nylon for strength and to reduce shadowing. Support ribs  52 , feed arm support rods  26 , and push rods  28  can be made of a fiberglass reinforced material, such as ¼-inch diameter G-10 rod for the support ribs and push rods for strength, and ⅛-inch diameter G-10 rod for the feed arm support rods. Center support rod  22  can comprise any conductive material having sufficient strength to support the antenna, but is preferably lightweight and machinable, and can be made from aluminum stock. In order for center feed cable  76  to be fed up through center support rod  22  as described above, center support rod  22  is preferably hollow. Center support rod  22  may also comprise other electrically conductive materials. 
     The reflector material is preferably strong and lightweight, and flexible enough to collapse. The electrical surface resistivity of the reflector is less than 0.5Ω/square, preferably less than 0.1Ω/square. The reflector is preferably made of a flexible conductive material, such as a copper and nickel plated rip-stop nylon, such as manufactured by ATM Flectron. The reflector is more preferably comprised of a conductive mesh fabric with a metal coating, such as a nickel/silver metal coating, for example that made by Swift Textile. The advantage of the reflector being comprised of a conductive mesh is reduced wind loading and improved dimensional stability which is particularly useful when the CIRA or CMIRA is deployed in the field. Alternatively, the reflector can be made of a metal-coated plastic film or a conductive mesh wire. A variety of types of conductive coatings can be used on the reflector material, such as nickel, copper, silver, gold, or brass. The feed arms preferably comprise a flexible, solid conductive material, such as conductive rip-stop nylon. The resistive loads on each feed arm preferably have an impedance in the range of 100 to 300Ω. The fabric resistors typically used for the resistive loads preferably have a surface resistivity in the range of 200Ω/square, such as can be achieved with polypyrrole treated woven polyester cut to form a 200Ω (±10%) resistor so that the TDR is not compromised, such as manufactured by Milliken Research Corp. 
     INDUSTRIAL APPLICABILITY 
     The invention is further illustrated by the following non-limiting examples. 
     EXAMPLE 
     Both the CIRA and CMIRA embodiments were tested using standard time domain antenna range techniques, and the results were converted to IEEE standard gain in the frequency domain. Two CIRA configurations were tested, an ultra-lightweight configuration having twelve triangular panels and a twenty-panel configuration. One CMIRA configuration was tested, having twenty panels, in both the focused and defocused modes. 
     Normalized Impulse Response 
     First, a review of the parameters used to describe antennas is provided. Antennas are described in the time domain with an impulse response, of the form h N (t). In transmission mode, the antenna radiates a field on boresight, E rad (t), which is described by equation (6.5) in E. G. Farr and C. E. Baum,  Time Domain Characterization of Antennas with TEM Feed , Sensor and Simulation Note 426, October 1998, the content of which is incorporated herein by reference:                    E   rad          (   t   )           Z   o         =       1     2                 π                 rc                           h   N          (   t   )                  ∘                1       Z   c                         V   src          (   t   )              t                 (   1   )                         
     where Z o  is the impedance of free space, Z c  is the impedance of the 50Ω feed cable, r is the distance out the observation point on boresight, V src (t) is the source voltage measured into a 50-ohm load, c is the speed of light in free space, and the “°” symbol indicates convolution. In reception mode the antenna is described by equation (7.5) in the  Time Domain Characterization , Note 426, article incorporated above:                    V   rec          (   t   )           Z   c         =         h   N          (   t   )                  ∘                      E   inc          (   t   )             Z   o                   (   2   )                         
     where E inc (t) is the incident electric field on boresight. Note that the normalized impulse response, h N (t), completely describes the behavior of antennas with transverse electromagnetic (TEM) feeds in both transmission and reception. With both a transmitting and receiving antenna, the received voltage can be related to the source voltage by combining the above two equations, equation (8.1) of the  Time Domain Characterization , Note 426, article:                  V   rec          (   t   )       =       1     2                 π                 rc                           h     N   ,   RX            (   t   )                  ∘                  h     N   ,   TX            (   t   )                  ∘                         V   src          (   t   )              t                   (   3   )                         
     where h N,RX (t), is the normalized impulse response of the receive antenna and h N,TX (t) is the response of the transmit antenna. 
     To calibrate the measurement system, two different TEM sensors are used. In this case, the antenna equation becomes:                  V   rec          (   t   )       =       1     2                 π                 rc                           h     N   ,   tem            (   t   )                  ∘                  h     N   ,   tem            (   t   )                  ∘                         V   src          (   t   )              t                   (   4   )                         
     which is very similar to equation (4.1) of the  Time Domain Characterization , Note 426, article. The normalized impulse response of the sensors can be extracted from Equation (4) above as equation (8.2) in the  Time Domain Characterization , Note 426, article:                    h   ~       N   ,   tem            (   ω   )       =         2                 π                 rc                       V   ~     rec          (   ω   )           j                 ω                       V   ~     src          (   ω   )                     (   5   )                         
     The details of this sensor calibration are included in the section entitled “IEEE Standard Gain” herein. Once a calibration has been performed with two identical antennas, then the response of an antenna under test is measured by replacing one of the sensors with the antenna under test. The impulse response of the antenna then becomes:                    h   ~       N   ,   AUT            (   ω   )       =       2                 π                 rc                       V   ~     rec          (   ω   )             j                 ω                       V   ~     src          (   ω   )              h     N   ,   tem            (   ω   )                                (   6   )                         
     and the time domain normalized impulse response is found with an inverse Fourier transform. 
     As a check on the reasonableness of the measurement, an aperture height, h a , is typically calculated which can be related to the physical parameters of the antenna under test. To find the aperture height it is necessary to convert the normalized impulse response to the conventional impulse response. This conversion is given by equation (7.4) of  Time Domain Characterization , Note 426:                  h     N   ,   RX            (   t   )       =         τ     p   ,   RX           f     g   ,   RX                             h   RX          (   t   )                 (   7   )                         
     where τ p,RX  is defined as:                τ     p   ,   RX       =       2            Z   c          Z     a   ,   RX                 Z   c     +     Z     a   ,   RX                   (   8   )                         
     and f g,RX  is defined as:                f     g   ,   RX       =       Z     a   ,   RX         Z   o               (   9   )                         
     Here, Z c  is the cable impedance (50Ω), Z a  is the antenna impedance, and Z o  is the impedance of free space (376.727Ω). Since all measurements taken have the antenna under test as the receiver, only the “RX” versions of the equations are included here. For the 100Ω TEM horn sensor used to make the antenna measurements, τ p,RX =0.942 and f g,RX =100/Z o =0.265. For the CIRA and CMIRA embodiments of the invention, which have splitters in the feed circuit, τ p,RX ≈1 (from section VII of  Time Domain Characterization , Note 426) and Z a =200Ω for one feed arm so f g,RX =200/Z o =0.531. The integral of the conventional impulse response is used later to determine the aperture height for both the sensor and the CIRA. The aperture height, h a , corresponds to the jump in the integral                h     a   ,   RX       =       ∫   Impulse              h   RX          (   t   )               t                 (   10   )                         
     The aperture height is useful since the effective height (at midband) relates the incident electric field to the voltage into a scope by a simple proportionality (equation (3.4) of  Time Domain Characterization , Note 426): 
     
       
           V   rec ( t )≈ h   eff   E   inc ( t )  (11) 
       
     
     where 
     
       
           h   eff =τ RX   h   a,RX   (12) 
       
     
     and                τ   RX     =       2                   Z   c           Z   c     +     Z     a   ,   RX                   (   13   )                         
     For the 100Ω TEM horn, τ RX =0.667 and for the CIRA and CMIRA τ RX =0.50. 
     IEEE Standard Gain 
     It is frequently desirable to convert the impulse response developed in the previous section to IEEE standard gain. The IEEE standard gain is more widely accepted as a measure of antenna performance than the normalized impulse response. The derivation of the conversion process is provided here. Here the IEEE gain is expressed in terms of the normalized impulse response, h N (t). 
     To begin, the standard expressions are provided in the frequency domain. Thus, the received power is: 
     
       
           P   rec   =A   eff   S   inc   (14) 
       
     
     where S inc  is the incident power density in Watts/m 2  and A eff  is the effective aperture. Gain is related to effective aperture by:                A   eff     =         λ   2       4                 π                     G             (   15   )                         
     Combining the above two equations:                P   rec     =           λ   2        G       4                 π                       S   inc               (   16   )                         
     Take the square root, and recast into voltages, to find:                    V   rec          (   ω   )           Z   c         =         λ          G        (   ω   )             2        π                             E   inc          (   ω   )           Z   o                   (   17   )                         
     where Z c  is the cable impedance, generally 50Ω and Z o  is the impedance of free space, 377 Ω. 
     To compare the above equation to the standard equation for reception, Equation (2) above is converted into the frequency domain, obtaining:                    V   rec          (   ω   )           Z   c         =         h   N          (   ω   )                           E   inc          (   ω   )           Z   o                   (   18   )                         
     where h N (ω) is the normalized antenna impulse response expressed in the frequency domain. The normalized impulse response, h N (t), is already known. To convert it to gain, Equations (17) and (18) are combined:                G        (   ω   )       =           4                 π       λ   2                                h   N          (   ω   )            2       =         4                 π                   f   2         c   2                                h   N          (   ω   )            2                 (   19   )                         
     This formula allowed the conversion of the measured time domain impulse response to IEEE gain, so that it is consistent with others in the field. It is to be noted that the above gain is not quite consistent with the IEEE standard because it does not include return loss, which is typically small for this class of antennas over the frequency range of interest. As used herein, an antenna is defined as operational when having greater than 0 dB of gain, as defined in Equation  19 , for a given frequency. 
     Data Acquisition System and Sensor Calibration 
     The characteristics of the antennas were measured using time domain techniques. This was done for two embodiments of the CIRA, a 20-panel and an ultra-lightweight CIRA, as well as for the CMIRA in both focused and defocused modes. The time domain data was processed to obtain the normalized impulse response as described above. Data was collected at 2.5° intervals in the H and E planes and converted to IEEE standard gain. The conversion from impulse response to IEEE standard gain was based on the derivation above. The impulse response characteristics, standard gain, and antenna patterns in the H and E planes are presented. 
     The data acquisition system and sensor calibration are now described. The antenna measurement configuration used is shown schematically in FIG.  15 . It included a Picosecond Pulse Labs (PSPL) 4015C Step Generator, which drives TEM sensor  206 . Two different sensors were used for taking measurements; 100Ω (the Farr Research, Inc. FRI-TEM-02-100) and 50Ω (the Farr Research, Inc. FRI-TEM-01-50). These two sensors were chosen because the antennas were designed in these examples to operate over the range between 80 MHz and 2 GHz, although a much broader bandwidth was achieved. The larger sensor was used in order to obtain the best possible low-frequency measurement, due to its greater sensitivity or h eff , and for its clear time, while the smaller sensor was used to ensure observation of the fastest possible full width half maximum (FWHM) out of the CIRA. See Table 1 below. Both of these sensors are essentially a half TEM horn mounted against a truncated ground plane. (See FIG. 26, which shows the FRI-TEM-01-100 sensor.) Returning to FIG. 15, remote pulser head  208  is shown at the sensor site. On the receive end, antenna under test  200  receives the signal, which is sampled by the SD24 sampling head through a 61 cm Goretex cable  202  connection and stored by the Tektronix 11801B Digital Sampling Oscilloscope (DSO). A two meter extender cable  204  was used between the sampling head and the DSO. The DSO communicated with the step generator on trigger line  209  to control the timing. Data was then downloaded to a computer for processing by way of a general purpose interface bus (GPIB) connection. The output of the PSPL 4015 C was a four volt step with a risetime of 20 picoseconds. 
     
       
         
           
               
             
               
                 TABLE 1 
               
             
            
               
                   
               
               
                 Characteristics of FRI-TEM Sensors 
               
            
           
           
               
               
               
               
               
               
            
               
                   
                   
                   
                   
                 3 dB 
                 Clear 
               
               
                   
                 Ground plane 
                 Impedance 
                 h eff ** 
                 point 
                 Time 
               
               
                 Model Number 
                 mm 
                 Ω 
                 mm 
                 GHz 
                 ns 
               
               
                   
               
               
                 FRI-TEM-01-50  
                 254 × 610  
                  50 
                 17 
                 12 
                 2 
               
               
                 FRI-TEM-01-100 
                 254 × 610  
                 100 
                 21 
                 10 
                 2 
               
               
                 FRI-TEM-02-50  
                 508 × 1220 
                  50 
                 30 
                  7* 
                 4 
               
               
                 FRI-TEM-02-100 
                 508 × 1220 
                 100 
                 42 
                  6 
                 4 
               
               
                   
               
               
                 *Estimated  
               
               
                 **V out (t) ≈ h eff  × E inc (t).  
               
            
           
         
       
     
     Calibration 
     The FRI-TEM-02-100 horn was a 100Ω sensor with a ground plane measuring 20×48 inches (508×1220 millimeters). The time domain reflectometry plot (TDR) of the sensor is shown in FIG. 16 a . The feed point is indicated on the response at  210  and the aperture is indicated at  212 . In order to calibrate the sensor, the antenna under test shown in FIG. 15 was replaced with a second identical FRI-TEM-02-100 sensor and the sensor was calibrated according to the techniques described in the  Time Domain Characterization of Antennas , Note 426, cited above. The calibration was performed with the sensor apertures 20 meters apart and 3.0 meters above the ground. This provided a delay of 3.0 nanoseconds before the ground bounce signal arrived. 
     The calibration of the FRI-TEM-02-100 sensor is provided in FIGS. 16 b - 16   d . The signal was truncated at the receiving sensor shortly after the impulse to remove the ground bounce. Also, the signal was zero-padded out to 20 ns for processing, to improve the low frequency response. The normalized impulse response had a FWHM of 50 ps as shown in FIG. 16 b . The frequency response was extremely flat as shown in FIGS. 16 c  and  16   d . This sensor had a clear time of 4 ns and a maximum gain of about  17   d B. The 100Ω impedance of this antenna increases the sensitivity, due to the increased effective height, but causes a mismatch into 50Ω cables. However, the improved sensitivity more than offset the effect of the mismatch. 
     Next, the FRI-TEM-01-50 sensor was calibrated. This sensor has a 50Ω impedance to match 50Ω cables. The ground plane for this sensor measures 10×24 inches (254×610 millimeters). The TDR of the sensor is shown in FIG. 17 a  and the feed point can be seen at  213 . As above, two identical FRI-TEM-01-50 sensors were used to calibrate the sensor. The calibration was performed with the sensor apertures 10 meters apart and 3.0 meters above the ground. This provided a 5.5 ns delay before the ground bounce signal arrived. 
     The calibration of the FRI-TEM-01-50 sensor is provided in FIGS. 17 b - 17   d . No truncation or zero padding of the signal was required. The normalized impulse response had a FWHM of 31 ps as shown in FIG. 17 b . The frequency response was extremely flat as shown in FIGS. 17 c  and  17   d . This sensor had a clear time of 2 ns and a maximum gain of approximately 15 dB. 
     Ultra-Lightweight CIRA Measurement Data 
     An ultra-lightweight CIRA was tested that was comprised of twelve triangular panels connected to a common center circular panel as shown and described above with respect to FIGS. 1-10. With twelve panels used for the CIRA reflector, the diameter of the center can that supported the antenna and contained the RF splitter was able to be reduced from that required by the 20-panel configuration described below. Fewer support ribs were also required for this configuration making it easier to deploy than the 20-panel configuration. The reflector was made of conductive mesh fabric having a silver and nickel plating with a resistance of less than 0.2Ω/square. The air permeability of the fabric was approximately 19.3 (m 3 /s)m 2  or 3800 (ft 3 /min.)ft 2  which provided greatly reduced wind loading. The conductive mesh reflector was also somewhat transparent, thereby reducing visibility of the antenna in the field to some extent. The four feed arms were made from copper and nickel plated conductive rip-stop nylon having an electrical surface resistivity of less than 0.1Ω/square. Rip-stop nylon was used for the feed arms as tests showed that this material provided a flatter TDR and better overall antenna performance than when the feed arms were made from the same conductive mesh fabric as the reflector. The resistive loads on each feed arm were made of polypyrrole treated woven polyester with a surface resistivity in the range of 200Ω/square. 
     When in the collapsed position, the ultra-lightweight CIRA measured 102 mm (4 inches) in diameter by 81 cm (32 inches) in length, and it weighed 2 kg, or 4.5 lbs. The reflector was 1.22 m (48 inches) in diameter with F/D=0.4 and had a depth of approximately 190 mm. The reflector frame comprised fiberglass support ribs connected to an aluminum center support rod by aluminum pivots. The splitter consisted of a 50Ω input impedance connector which split into two 95Ω cables that attached to the feed arms at the feed point in a series/parallel configuration in the standard IRA configuration having four feed arms. 
     This ultra-lightweight configuration had less aperture blockage than the 20-panel CIRA configuration discussed below due to its smaller number of push rods. The variation of the reflector from the desired paraboloid for the ultra-lightweight configuration was approximately ±10 mm as measured from the focal point. It was found that too much variation in the shape of the reflector caused severe degradation of the impulses response and beam shape. This was demonstrated by the 20-panel CIRA and CMIRA configurations described below, the reflectors of which were constructed too flat, causing them to be somewhat out of focus. However, this can be explained by the stretch in the rip-stop nylon fabric used for the reflectors of each of those configurations, as well as by small variations in the cutting and sewing of the panels. The ultra-lightweight configuration had an improved response due in large part to the reduced stretch of the tough conductive mesh used for the reflector, improved fabric patterns, sewing techniques, and greater quality control. The test data presented herein will be understood by those skilled in the art not to limit the scope of the invention but instead to demonstrate the capabilities of but a few possible configurations of the invention based upon the basic principles for a collapsible IRA set forth herein. 
     The characteristics of the ultra-lightweight CIRA were measured using the available time domain outdoor antenna range of Farr Research, Inc. Both the FRI-TEM-02-100 and FRI-TEM-01-50 horn sensors were used for these measurements, and the antenna was measured with the data acquisition system shown in FIG.  15 . The distance between the antennas was twenty meters and the height was three meters above the ground. Antenna pattern measurements in the H and E planes were made at 2.5° increments. Also, the IEEE standard gain was computed, and plotted on boresight as a function of frequency and at various frequencies as a function of angle in the principal planes. 
     The TDR of the antenna is shown in FIG. 18 a . The connector can be seen at  214 , the splitter at  216 , the feed cable in the area shown by  218 , the feed point at  220 , and resistors at  222 . The TDR at the feed point and the feed arms was very good for four feed arm IRAs. In FIGS. 18 b ,  18   d , and  18   f  the on-boresight characteristics of the CIRA are shown as measured using the FRI-TEM-02-100 sensor. FIGS. 18 c ,  18   e , and  18   g  show the same measurements using the FRI-TEM-01-50 sensor. For the larger FRI-TEM-02-100 sensor the data were clipped just before the arrival of the ground bounce signal and then zero padded out to 20 ns, to provide frequency information down to 50 MHz. No modifications were made to the data from the small FRI-TEM-01-50 sensor. The measurements using the two sensors were almost identical. The FWHM of the normalized impulse response, shown in FIGS. 18 d  and  18   e , was 73 ps when measured with the larger sensor and slightly smaller (68 ps) when measured with the smaller sensor. The CIRA proved to be usable from below 50 MHz to above 8 GHz, as shown in FIGS. 18 f ,  18   g ,  18   h , and  18   i.    
     When deciding the distance at which to place the sensor, it must be taken into account that the far-field begins at a distance that is dependent upon the smallest FWHM expected to be measured. A FWHM of around 100 ps was expected to be measured, so a distance of 20 meters was chosen as adequate. However, with the 70 ns FWHM measurements, this faster impulse width extended the far field to around 25 m, using the formula r&gt;(3/2) a 2 /(ct FWHM ), where a is the antenna radius, c is the speed of light in free space, and t FWHM  is the FWHM of the radiated impulse response. While there was no opportunity to make new measurements at a greater distance, the error in the measurement was believed to be small. 
     Next, the gain vs. frequency is shown in FIGS. 18 h  and  18   i . The high-frequency response is approximately smooth to 8 GHz. The peak gain was 23 dB at 4 GHz. The h a  of the antenna from the integral of the impulse response (see FIGS. 18 j - 18   m ) was 0.30 m, so the midband effective height of the antenna was 15 cm. This value for h a  is 76% of the theoretical value of 0.396 m given in the  Time Domain Characterization , Note 426, article cited above, and in L. H. Bowen, E. G. Farr, and W. D. Prather,  Fabrication and Testing of Two Collapsible Impulse Radiating Antennas , Sensor and Simulation Note 440, November 1999. 
     In FIG. 19 a  the cross polarization (crosspol) response of the ultra-lightweight CIRA is shown. The IEEE gain on boresight for the crosspol case is shown in FIG. 19 b . The crosspol response is 10-20 dB below the coplanar polarization (copol) response from FIGS. 18 h  and  18   i . This data is of interest due to recent work suggesting improvements in the IRA that would result in improved gain and reduced crosspol. (See J. S. Tyo,  Optimization of the Feed Impedance for an Arbftrary Crossed-Feed-Arm Impulse Radiating Antenna , Sensor and Simulation Note 438, November 1999.) This is accomplished placing the feed arms at ±30 degrees from the dominant polarization angle, instead of ±45 degrees, as shown in FIG.  3 . Since each panel of the CIRA is 30 degrees wide, the feed arm positions can be configured at ±30 degrees as would be apparent to those skilled in the art. 
     In FIGS. 20 a  through  20   h , plots of the principal plane pattern cuts of the antenna at various frequencies from 98 MHz to 4,004 MHz as a function of angle off boresight in the H plane are shown. FIGS. 21 a  through  21   h  plot the principal plane pattern cuts of the antenna at various frequencies from 98 MHz to 4,004 MHz as a function of angle off boresight in the E plane. The data in each of these figures was measured and acquired using the system of FIG. 15 with the FRI-TEM-02-100 sensor. Turning to FIGS. 22 through 25, the antenna pattern in the H and E planes, based on the peaks of the raw voltage measurements are shown. As the antenna was turned on the tripod, the peak field tended to shift in time with the tripod mounting, and no attempt was made to adjust the time delay in the raw data to compensate for this. The half voltage beamwidth was 5.1° in the H plane and 6° in the E plane. The half power beamwidth was ˜3° in both the H and E planes. 
     20-Panel CIRA Measurement Data 
     Similar measurement data was taken for a 20-panel configuration of the CIRA using the data acquisition system of FIG. 15 except that only the FRI-TEM-02-100 sensor was used, that sensor being an earlier model than that used for the ultra-lightweight CIRA measurements above. This CIRA contained twenty of the triangular panels, was approximately 165 mm in depth, and was based, upon a 1.22 meter (48 inch) diameter parabolic dish with a focal length of 0.488 meters (F/D=0.4). The deviation from the ideal depth of 190 mm was due to stretch and wrinkles in the rip-stop nylon and inaccuracies in sewing, as discussed above. The panels making up the reflector as well as the feed arms were made from conductive rip-stop nylon having an electrical surface resistivity of less than 0.1Ω/square. This material was both strong and lightweight. The resistive loads on the feed arms were constructed of polypyrrole treated woven polyester with a surface resistivity in the range of 200 Ω/square. 
     This configuration was slightly over 127 mm (5 inches) in diameter and 737 mm (29 inches) long in the collapsed position and weighed approximately 2.8 kg (6 lb.). The splitter consisted of a 50Ω input impedance connector, which then split into two 95Ω cables. 
     The TDR of the FRI-TEM-02-100 sensor used in the data acquisition system when measuring the 20-panel CIRA configuration is shown in FIG. 30 a . The feed point can be seen at  224 , the support posts at  226 , and the aperture at  228 . In order to calibrate the sensor, the antenna under test was replaced with a second identical FRI-TEM-02-100 sensor. The calibration was performed with the sensor apertures 20 m apart and 2.1 m above the ground. This provided 1.5 ns delay before the ground bounce signal arrived. 
     The sensor calibration data is presented in FIGS. 30 b  and  30   c . The voltage measured at the receiving sensor was truncated shortly after the impulse signal to remove the ground bounce. Also, the signal was zero padded out to 20 ns for processing to improve the low frequency response. FIG. 30 b  provides the normalized impulse response of the sensor. The frequency response was extremely flat as shown in FIG. 30 c . There was a jump in the integral of the conventional impulse response which gave a value for h a  of 62.5 mm. The aperture height was 125 mm which gives a theoretical value for h a  of 62.5 mm. Therefore, the measured value was equal to the expected value. The effective height at midband, h eff =h a ×τ RX =40 mm, since τ RX =0.667. 
     When measuring the 20-panel configuration of the CIRA, the distance between the antennas was 20 m and the height was 3 m. The antenna was mounted on a tripod for testing. Antenna patterns in the H-and E-planes were made at 2.5° increments. The data were zero-padded out to 20 ns to provide information on the frequency response down to 50 MHz. Also, the IEEE standard gain was computed and plotted on boresight as a function of frequency. 
     The observed data were as follows. FIGS. 31 a  through  31   c  provide the boresight characteristics of the 20-panel CIRA. The FWHM of the normalized impulse response is 105 ps as shown in FIG. 31 a . FIG. 31 b  shows the normalized impulse response and FIG. 31 c  is the IEEE gain as a function of frequency on boresight. The antenna was usable from below 50 MHz to above 8 GHz. The midband effective height of the antenna was found from the integral of the impulse response to be 0.31 m. This was 78% of the theoretical value of 0.396 m. There were some bumps in the boresight impulse response of the 20-panel CIRA at low frequencies, as shown in FIG. 31 b . Attempts were made to reduce these bumps by eliminating the ground reflection. With the ground reflection eliminated, the low-frequency frequency bumps in the resulting impulse response were indeed smoother, but they were not eliminated. 
     The antenna pattern in the H plane, based on the peaks of the raw voltage measurements, is shown in FIG.  32 . FIG. 33 contains similar data for the E plane. The half voltage beamwidth is 13° in the H plane and 11° in the E plane. The half power beamwidths are 7° and 6° in the H and E planes respectively. 
     Based on this data, the beam width was considered. In E. G. Farr, C. E. Baum, and W. D. Prather,  Multifunction Impulse Radiating Antennas: Theory and Experiment , Sensor and Simulation Note 413, November 1997, the half field beamwidth (HFBW) is defined as the angle between the two locations in a pattern cut where the field is down by half from the peak. Since the measured (raw) voltage is proportional to the incident electric field, this is the same as the half voltage beamwidth used above. Using this definition and the calculation methods of Simulation Note 413 cited above, the HFBW in the H plane can be estimated to be 3° and in the E plane to be 4° for an ideal antenna. The theoretical fields at discrete angles of 0°, 1°, 2°, and 5° off boresight were used for the above estimates. The angles for the 20-panel CIRA were 3.5-4.6 times these values. This is due in large part to the antenna being somewhat out of focus due to the curvature of the reflector, stretch of the fabric, and sewing, as discussed above. 
     CMIRA Measurement Data 
     A 20-panel CMIRA configuration was also tested in both the focused and defocused modes using the same 100Ω TEM horn as described above and used in measuring the 20-panel CIRA. This embodiment had four expansion seams in the reflector, as shown in FIG. 13, in four places near the perimeter of the reflector and spaced apart at 90 degree intervals. In the focused mode, these seams were held in the folded position by means of conductive mating Velcro® connectors. In the defocused mode the reflector was flattened by releasing the mating Velcro® connectors, allowing the expansion seams to unfold and conduct. Extension sections were placed in the feed arms between the resistive loads and the reflector to enable the defocused mode as described with respect to FIG.  14 . 
     As with the 20-panel CIRA, the reflector for the 20-panel CMIRA that was tested was made from conductive copper and nickel plated rip-stop nylon, as were the feed arms. The resistive load on the feed arms was made from polypyrrole treated woven polyester. The splitter consisted of a 50Ω input impedance connector, which split into two 95Ω cables. 
     The 20-panel CMIRA to be tested was designed to have a diameter of 1.22 m (48 inches) and a focus of 0.488 m (19.2 inches) in the focused mode. This would provide a ratio F/D of 0.40 and a depth of 190 mm (7.5 inches) in the focused mode. However, as with the 20-panel CIRA tested, the stretch of the rip-stop nylon reflector and slight variations in sewing the reflector panels together caused some deviations from an ideal parabolic reflector dish. Therefore, the depth of the CMIRA in the focused mode was approximately 146 mm rather than the ideal 190 mm. 
     The data for the focused CMIRA are shown in FIGS. 34-36. FIGS. 34 a-c  provide the boresight characteristics of the focused CMIRA. FIG. 34 c  is the IEEE gain vs. frequency. The antenna is usable from below 100 MHz to above 7.5 GHz in the focused mode. The gain of the CMIRA at higher frequencies was somewhat lower than the gain of the 20-panel CIRA. Even in the focused mode the CMIRA was much flatter than ideal. This was due to stretch in the rip stop nylon reflector and in the Velcro® used to adjust the surface curvature of the reflector. Therefore, the data from the 20-panel CIRA presented above are expected to be more typical of the focused CMIRA than the data presented here. The midband effective height of the antenna was found from the integral of the impulse response to be 0.26 m. This height was expected to be the same as the CIRA. 
     As mentioned above, the CMIRA in the focused mode was out of focus by approximately 44 mm. This is 19 mm more than the 20-panel CIRA. Therefore, the differences in the responses of the CIRA and CMIRA were primarily a result of the difference in reflector depths, not the presence of the expansion seams in the CMIRA. Because of this, the effects due to the expansion seams alone were unable to be isolated. 
     The antenna pattern in the H plane, based on the peaks of raw voltage measurements, is shown in FIG.  35 . FIG. 36 contains similar data for the E plane. From the peak values at various angles, the beamwidth in the major planes can be determined. The half voltage beamwidth was 24° in the H plane and 17° in the E Plane. The beamwidth for the 20-panel CIRA was at least 6° narrower in both planes. For the half power case, there are beamwidths of 10° in the H plane and 11° in the E plane. 
     The same measurements as above were taken for the CMIRA in the defocused mode, and the data is provided in FIGS. 37-39. The IEEE gain as a function of frequency is shown in FIG. 37 c . From FIG. 37 c  it can be observed that the defocused CMIRA had a low-end 3 dB frequency of around 100 MHz and a high-end 3 dB frequency of around 1 GHz. The midband effective height of the antenna was found from the integral of the impulse response to be 0.21 m, an interesting result, since the midband effective height for the defocused configuration should have been the same as that for the focused configuration, or 0.32 m. 
     The antenna pattern in the H plane, based on the peaks of raw voltage measurements, is shown in FIG.  38 . FIG. 39 provides similar data for the E plane. The half voltage beamwidth was 76° in the H plane and 32° in the E plane. The beamwidth was much wider for the defocused case, which was as expected, being the reason for building a defocused antenna. The half power beamwidths were 68° and 20° in the H and E planes respectively. 
     The preceding examples can be repeated with similar success by substituting the generically or specifically described operating conditions of this invention for those used in the preceding examples. 
     Although the invention has been described in detail with particular reference to these preferred embodiments, other embodiments can achieve the same results. Variations and modifications of the present invention will be obvious to those skilled in the art and it is intended to cover in the appended claims all such modifications and equivalents. The entire disclosures of all references, applications, patents, and publications cited above are hereby incorporated by reference.