Patent Publication Number: US-9900004-B2

Title: Radio frequency switching circuit with distributed switches

Description:
CROSS REFERENCE TO RELATED APPLICATION—CLAIM OF PRIORITY 
     This application is a continuation of commonly owned and co-pending U.S. patent application Ser. No. 14/995,023 filed Jan. 13, 2016, entitled “Radio Frequency Switching Circuit with Distributed Switches”, the disclosure of which is incorporated herein by reference in its entirety. Application Ser. No. 14/995,023 is a continuation-in-part of commonly owned and co-pending U.S. patent application Ser. No. 14/610,588 filed Jan. 30, 2015, entitled “Radio Frequency Switching Circuit with Distributed Switches”, the disclosure of which is incorporated herein by reference in its entirety. 
    
    
     BACKGROUND 
     (1) Technical Field 
     This invention generally relates to electronic signal switching devices, and more specifically to electronic radio frequency signal switching devices. 
     (2) Background 
     Electronic signal switches are used in a wide variety of applications. One type of signal switch in common use is a field effect transistor (FET) that is actively controlled through a gate terminal to block or pass an electrical signal connected in series with source and drain terminals of the FET (in another mode of operation, a FET also may be used to modulate an electrical signal in response to a varying signal on the gate terminal). 
     Field effect transistors may be fabricated in various technologies (e.g., standard bulk silicon, silicon-on-insulator, silicon-on-sapphire, GaN HEMT, GaAs pHEMT, and MESFET processes) and are commonly represented in schematic diagrams as an idealized device. However, in many applications, particularly in radio frequency (RF) circuits, the structure and materials of a FET switch may have significant effects on its own operation (e.g., with respect to bandwidth, isolation, and power handling) and the presence of a FET switch may have significant effects on other components in a circuit. Such effects arise in part because a “CLOSED”/“ON” (low impedance) FET has a non-zero resistance, and an “OPEN”/“OFF” (high impedance) FET behaves as a capacitor due to parasitic capacitances arising from the proximity of various semiconductor structures, particularly within the close confines of an integrated circuit (IC). Large signal behaviors affecting power handling may also arise from other characteristics of a FET, such as avalanche breakdown, current leakage, accumulated charges, etc. Accordingly, the actual in-circuit behavior of a FET must be taken into account when designing FET based circuitry. 
     One use of FET switches is within RF frequency signal switching devices. For example,  FIG. 1A  is a schematic diagram of a prior art 3-port reflective signal switching device  100  for selectively coupling one of two terminal ports  102 A,  102 B (shown series connected to respective external loads RF 1 , RF 2 ) to a common port  104  (shown series connected to an external load RFC). Accordingly, the signal switching device  100  may be regarded as a single-pole, double-throw (SPDT) switch. In other configurations, more than two terminal ports (a 1×N switch) and more than one common port may be included (an M×N switch). Between the common port  104  and each terminal port  102 A,  102 B are respective FET series switches  106 A,  106 B; the FET series switches  106 A,  106 B may vary in size, for example, to accommodate different power levels. Between each terminal port  102 A,  102 B and its respective series switch  106 A,  106 B are respective FET shunt switches  108 A,  108 B, coupled to circuit ground. Such a switching device  100  may be used, for example, to selectively couple RF signals between two antennas respectively connected to the terminal ports  102 A,  102 B and transmit and/or receive circuitry connected to the common port  104 . For RF signals, each load/source impedance RF 1 , RF 2 , RFC would typically have a nominal impedance of 50 ohms by convention. 
     In operation, when terminal port  102 A is to be coupled to the common port  104 , series switch  106 A is set to a low impedance ON state by means of control circuitry (not shown) coupled to the gate of the FET series switch  106 A. Concurrently, shunt switch  108 A is set to a high impedance OFF state. In this state, signals can pass between terminal port  102 A and the common port  104 . 
     For the other terminal port  102 B, the series switch  106 B is set to a high impedance OFF state to decouple the terminal port  102 B from the common port  104 , and the corresponding shunt switch  108 B is set to a low impedance ON state. One purpose of setting the shunt switch  108 B to ON—thus coupling the associated terminal port  102 B to circuit ground—is to improve the isolation of the associated terminal port  102 B (and coupled circuit elements, such as antennas) through the corresponding series switch  106 B. For switching devices with more than two terminal ports, the series switch and shunt switch settings for the “unused” (decoupled) terminal port to common port signal paths typically would be set to similar states. 
       FIG. 1B  is a diagram showing an equivalent circuit model of the prior art 3-port signal switching device of  FIG. 1A . Shown is a circuit configuration  120  in which terminal port  102 A has been coupled to the common port  104 ; accordingly, series switch  106 A and shunt switch  108 B are set to a low impedance ON state, while series switch  106 B and shunt switch  108 A are set to a high impedance OFF state. In this configuration, series switch  106 A is modeled as a resistor  126 A having a resistance value of Ron (i.e., the CLOSED or ON state resistance of a FET), shunt switch  108 A is modeled as a capacitor  128 A having a capacitance of Cshunt (i.e., the OPEN or OFF state capacitance of a FET), series switch  106 B is modeled as a capacitor  126 B having a capacitance of Coff, and shunt switch  108 B is modeled as a resistor  128 B having a resistance value of Rshunt. As in  FIG. 1A , with the illustrated circuit configuration, signals can pass between terminal port  102 A and the common port  104 . 
       FIG. 1C  is a diagram showing a simplified equivalent circuit model  130  corresponding to the circuit configuration  120  shown in  FIG. 1B . Series switch  106 B (modeled as a capacitor  126 B in  FIG. 1B ) is OFF. The corresponding shunt switch  108 B (modeled as a resistor  128 B in  FIG. 1B ) is ON, thus having a very low impedance and coupling terminal port  102 B to circuit ground. Since Rshunt has a very low impedance, the resistor equivalent  128 B in  FIG. 1B  may be more simply modeled as a conductor (short) to circuit ground and is thus shown in dotted-line resistor form. Therefore, the two equivalent circuit elements  126 B,  128 B of  FIG. 1B  may be modeled as a single capacitor  126 B′ having a capacitance of Coff. Similarly, since series switch  106 A (modeled as a resistor  126 A in  FIG. 1B ) is ON and Ron is a very low impedance, series switch  106 A may be more simply modeled as a conductor. Accordingly, the resistor equivalent  126 A in  FIG. 1B  is shown in dotted-line resistor form, leaving OFF shunt switch  108 A (modeled as a capacitor  128 A with a capacitance of Cshunt) connected in parallel with the external load RF 1 . As in  FIG. 1A  and  FIG. 1B , with the illustrated circuit configuration, signals can pass between terminal port  102 A and the common port  104 , as shown by dotted line signal path  132 . 
     The simplified equivalent circuit model  130  can be used to evaluate the insertion loss (IL) bandwidth of the circuit model  130 . In this example, the 3 dB IL bandwidth is proportional to 1/(Rport*(Coff+Cshunt)) [where Rport is the load resistance at the RF 1  and RFC ports], which is typically limited to below 13 GHz in current silicon IC technology. 
     The bandwidth of conventional radio frequency switching devices of the type shown in  FIGS. 1A, 1B, and 1C  is limited by the parasitic capacitance from the Cshunt equivalent components. This invention in various embodiments addresses this limitation to improve the bandwidth of RF switching devices as well as the signal isolation and power handling of such switching devices. 
     SUMMARY OF THE INVENTION 
     Embodiments of the invention use distributed shunt switches distributed along transmission lines (or may include other inductive impedance compensating components) to improve RF bandwidth with respect to insertion loss, and to improve isolation. In addition, the shunt switches may be physically positioned on both sides of the transmission lines to keep an integrated circuit (IC) design essentially symmetrical so as to provide predictable and reliable operational characteristics. Some embodiments include stacked FET shunt switches and series switches to tolerate high voltages. In some embodiments, the gate resistor for each FET shunt switch is divided into two or more portions to save IC area near the transmission lines, or to optimize a performance parameter, such as power handling, isolation, or low frequency behavior. 
     The details of one or more embodiments of the invention are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the invention will be apparent from the description and drawings, and from the claims. 
    
    
     
       DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  is a schematic diagram of a prior art 3-port reflective signal switching device for selectively coupling one of two terminal ports to a common port. 
         FIG. 1B  is a diagram showing an equivalent circuit model of the prior art 3-port signal switching device of  FIG. 1A . 
         FIG. 1C  is a diagram showing a simplified equivalent circuit model corresponding to the circuit configuration shown in  FIG. 1B . 
         FIG. 2A  is a schematic diagram of a 3-port signal switching device for selectively coupling one of two terminal ports to a common port in accordance with the teachings of this disclosure. 
         FIG. 2B  is a schematic representation of an elemental length of a transmission line. 
         FIG. 2C  is a diagram showing an equivalent circuit model of the 3-port signal switching device of  FIG. 2A . 
         FIG. 3  is a graph showing simulation results of three variations of a switching device in accordance with  FIG. 2A . 
         FIG. 4A  is a schematic diagram of a circuit architecture having distributed stacked shunt switches as well as distributed gate resistors. 
         FIG. 4B  is a schematic diagram of a circuit architecture having lumped stacked shunt switches. 
         FIG. 5  is a schematic diagram of a circuit architecture having stacked series switches. 
         FIG. 6  is a diagram of a conceptual circuit layout of an RF switching circuit with distributed stacked switches for both the shunt and series switch components. 
     
    
    
     Like reference numbers and designations in the various drawings indicate like elements. 
     DETAILED DESCRIPTION OF THE INVENTION 
     The bandwidth of conventional radio frequency (RF) switching devices of the type shown in  FIGS. 1A, 1B, and 1C  is limited by the parasitic capacitance from the Cshunt equivalent components. Embodiments of the invention use distributed shunt switches distributed along transmission lines (or may include other inductive impedance compensating components) to improve RF bandwidth with respect to insertion loss, and to improve isolation. In addition, the shunt switches may be physically positioned on both sides of the transmission lines to keep an integrated circuit (IC) design essentially symmetrical so as to provide predictable and reliable operational characteristics. Some embodiments include stacked FET shunt switches and series switches to tolerate high voltages. In some embodiments, the gate resistor for each FET shunt switch is divided into two or more portions to save IC area near the transmission lines, or to optimize a performance parameter, such as power handling, isolation, or low frequency behavior. 
     Distributed Shunt Switches 
       FIG. 2A  is a schematic diagram of a 3-port signal switching device  200  for selectively coupling one of two terminal ports  102 A,  102 B (shown series connected to respective external loads RF 1 , RF 2 ) to a common port  104  (shown series connected to an external load RFC) in accordance with the teachings of this disclosure. Accordingly, the illustrated signal switching device  200  may be regarded as a single-pole, double-throw (SPDT) switch. 
     Between the common port  104  and each terminal port  102 A,  102 B are respective primary isolation FET series switches  201 A,  201 B that operate in essentially the same fashion as the corresponding series switches  106 A,  106 B in  FIG. 1A . In other configurations, only one terminal port may be used (e.g., only terminal port  102 A in a single-pole, single-throw switch), more than two terminal ports may be included (e.g., a 1×N switch), and more than one common port may be included (e.g., an M×N matrix switch). When three or more terminal ports are included (e.g.,  102 A,  102 B, and a similar third port not shown), the switching device  200  may be utilized as a transfer switch, allowing a signal to be communicated from any one port to any other port while isolating unused ports from the signal path (the common port  104  is not used in such a case unless an isolation serial switch to the signal path is interposed). Further, the FET series switches  201 A,  201 B may be of different sizes in some embodiments. The illustrated embodiment may be advantageously embodied on a silicon-on-insulator (SOI) integrated circuit (IC) substrate. 
     An important aspect of the disclosed embodiments is that inductive tuning components are included to compensate for the OFF state capacitance Cshunt of the shunt switch units  204  described below. One way to provide such inductive tuning components is to use a transmission line that includes at least one series inductive component coupled to at least one shunt capacitive component. In the embodiment illustrated in  FIG. 2A , each of the FET series switches  201 A,  201 B is coupled to a corresponding transmission line  202 A,  202 B that can be modeled as a plurality of series-coupled inductive tuning components  203  (depicted as rectangular symbols in this example). Each transmission line  202 A,  202 B may be implemented, for example, as microstrips or coplanar waveguides. For RF switching devices, the transmission lines  202 A,  202 B would typically be tuned to have a nominal impedance of 50 ohms by convention. The individual shunt switches  108 A,  108 B shown in  FIG. 1A  have each been replaced by sets of n (where n≧1) parallel FET shunt switch units  204 . The shunt switch units  204  may be reduced in size compared to a conventional single shunt switch  108 A,  108 B. 
       FIG. 2B  is a schematic representation of an elemental length of a transmission line, where Rdx, Ldx, Gdx, and Cdx, are respectively, the per unit length resistance, inductance, conductance, and capacitance of the line. The impedance Zo of such a transmission line is Zo=√(Ldx/Cdx). The OFF state capacitance Cshunt of the shunt switches units  204  is in parallel with Cdx. To achieve compensation of Cshunt of the shunt switches units  204 , Ldx can be increased, or Cdx can be decreased, or both, with respect to each other, so that √(Ldx/(Cdx+Cshunt))=Zo (commonly specified as 50 ohms by convention). 
     In the illustrated embodiment, the conduction (source-drain) channel of each FET shunt switch unit  204  is coupled to circuit ground and between a corresponding pair of inductive tuning components  203 , thereby forming an elemental length of a transmission line  206 , examples of which are shown bounded by dotted boxes. In some embodiments, an inductive tuning component  203  may be shared between adjacent shunt switch units  204 , thus constituting part of two elemental lengths of a transmission line. However, for purposes of circuit analysis, it may be easier to model a shared inductive tuning component  203  as being “split” between adjacent shunt switch units  204 . 
     As more fully explained below, the primary series switches  201 A,  201 B and the shunt switch units  204  may be replaced by multiple series-coupled FET switches to tolerate higher voltages than a single FET switch. Such “stacking” of FET switches helps decrease the effective Cshunt while permitting higher power handling. 
     Additional supplemental inductive tuning components  207  (also labeled L a  and L b ) may be added at either end or both ends of the transmission lines  202 A,  202 B to enable fine tuning of parasitics unrelated to the transmission lines  202 A,  202 B, such as the series switch device parasitic capacitances and pad capacitance for I/O interconnects. The values for the supplemental inductive tuning components  207  (L a , L b ) of one transmission line may be the same or different with respect to each other, and with respect to the supplemental inductive tuning components  207  (L a , L b ) of other transmissions lines. In addition, in some applications, it may be useful to insert an additional secondary isolation series FET switch in series with the transmission line signal path at each node marked “X” in  FIG. 2A  and  FIG. 2C , making the circuit appear more symmetrical electrically. By opening the primary and secondary series FET switches at both ends of a transmission line (e.g.,  202 B) associated with an unused terminal port (e.g.,  102 B), that transmission line will be fully isolated from any circuitry attached to such ends. The secondary isolation series FET switches may be the same size as the primary isolation series switches  201 A,  201 B, but equal sizing is not required to still provide benefit. For example, modeling has shown that there may be a benefit to having different sizes for the secondary isolation series FET switches relative to the primary isolation series FET switches. In general, adding secondary isolation series FET switches would most frequently apply to switching devices having two or more terminal ports, but may also be applied to switching devices having a single terminal port (e.g., a SPST switch) when there may be some aspect of the external circuit element being switched where having more isolation on both sides of the transmission line signal path is helpful (e.g., when using an SPST embodiment in a shunt configuration to circuit ground). 
     In operation, when terminal port  102 A is to be coupled to the common port  104 , series switch  201 A is set to a low impedance ON state by means of control circuitry (not shown) coupled to the gate of the FET series switch  201 A. Concurrently, the set of n shunt switch units  204  coupled to transmission line  202 A is set to a high impedance OFF state. In this state, signals can pass between terminal port  102 A and the common port  104  along transmission line  202 A. 
     For the other terminal port  102 B in this example, the series switch  201 B is set to a high impedance OFF state to decouple transmission line  202 B and the terminal port  102 B from the common port  104 , and the set of n corresponding shunt switch units  204  coupled to transmission line  202 B is set to a low impedance ON state, thus coupling the associated terminal port  102 B to circuit ground. 
       FIG. 2C  is a diagram showing an equivalent circuit model of the 3-port signal switching device of  FIG. 2A . Shown is a circuit configuration  250  in which terminal port  102 A has been coupled to the common port  104 , as described with respect to  FIG. 2A . In this configuration, series switch  201 A is modeled as a resistor  210  having a resistance value of Ron, and series switch  201 B is modeled as a capacitor  212  having a capacitance of Coff. The shunt switch units  204  coupled to transmission line  202 A are each shown modeled as capacitances  220  with a capacitance of Cshunt/n. The shunt switch units  204  coupled to transmission line  202 B are each shown modeled as resistances  222  with a resistance of Rshunt*n. 
     As in  FIG. 2A , with the illustrated circuit configuration  250 , signals can pass between terminal port  102 A and the common port  104 . For RF signals, each load RF 1 , RF 2 , RFC would typically have a nominal impedance of 50 ohms by convention. Each of the inductive tuning components  203 ,  207  in  FIG. 2A  has a corresponding inductance, L a , L b , or L 1 , as shown in  FIG. 2C  and described below. Values for the inductive tuning components  203  may be selected to achieve compensation of Cshunt of the shunt switches units  204  as described above. Values for the supplemental inductive tuning components  207  (L a , L b ) may be selected to achieve compensation of Coff and the impedance (Zoff a , described below) of the OFF signal path, and for parasitics associated with signal interconnections. 
     Benefits of the embodiment illustrated in  FIG. 2A  and  FIG. 2C  include (1) tuning out the effect of Coff; (2) tuning out the effect of C shunt; and (3) improving isolation of OFF paths. 
     (1) Tuning Out the Effect of Coff 
     For the configuration shown in  FIG. 2A , the impedance Zoff of the OFF path (i.e., all of the elements from series switch  201 B through load RF 2 , as indicated by the dotted “Zoff” line in  FIG. 2C ) is given by the following formula:
 
 Z off= j ω( L   a   +L 1)+1/ jωC off+ Z off a   [Eq. 1]
 
where Zoff a  comprises the impedance of the OFF path after the first L 1  inductive tuning component through load RF 2  (as indicated by the dotted “Zoff a ” line in  FIG. 2C ) and whose value approaches n*Rshunt at higher frequencies.
 
     The resonant frequency of the Zoff impedance is 
               1     2   ⁢   π   ⁢         (       L   a     +     L   1       )     ⁢             ⁢             ⁢   Coff           .         
When Zoff is below its resonant frequency
 
               (       i   .   e   .     ,       jw   ⁡     (       L   a     +     L   1       )       &lt;          1     j   ⁢           ⁢   ω   ⁢           ⁢   Coff                )     ,         
achieved by selection of the values for the inductive tuning components  203  for a particular application, then the loading effect of the Coff capacitance on the ON path (i.e., all of the elements from series switch  201 A through load RF 1 ) is appreciably reduced, thus improving the bandwidth of the switching device  200  compared with conventional designs. This characteristic can be used to improve the design trade-off between bandwidth, insertion loss, and isolation for all such switching devices.
 
     (2) Tuning Out the Effect of Cshunt 
     For the configuration shown in  FIG. 2A , with respect to the ON path, the impedance Zon for the ON path (as indicated by the dotted “Zon” line) is given by the following formula: 
     
       
         
           
             
               
                 
                   Zon 
                   = 
                   
                     
                       
                         2 
                         * 
                         
                           L 
                           1 
                         
                       
                       
                         Cshunt 
                         / 
                         n 
                       
                     
                   
                 
               
               
                 
                   [ 
                   
                     Eq 
                     . 
                     
                         
                     
                     ⁢ 
                     2 
                   
                   ] 
                 
               
             
           
         
       
     
     The cutoff frequency (half power point), f c , is given by the following formula: 
     
       
         
           
             
               
                 
                   
                     f 
                     c 
                   
                   = 
                   
                     1 
                     
                       2 
                       ⁢ 
                       π 
                       ⁢ 
                       
                         
                           2 
                           * 
                           
                             L 
                             1 
                           
                           * 
                           
                             Cshunt 
                             n 
                           
                         
                       
                     
                   
                 
               
               
                 
                   [ 
                   
                     Eq 
                     . 
                     
                         
                     
                     ⁢ 
                     3 
                   
                   ] 
                 
               
             
           
         
       
     
     Accordingly, the half power point (3 dB) bandwidth of Zon is related to L 1 , n, and Cshunt, and can be adjusted by adding additional tuning network stages  206  (i.e., increasing n). The corresponding value of L 1  is then determined by Eq. 2 to maintain a constant Zon. As deduced from Eq. 2, as n is increased, the corresponding value of L 1  is decreased proportional to 1/n. In particular, the higher the number n of tuning networks  206 , the higher the cutoff frequency. For example,  FIG. 3  is a graph showing simulation results of three variations of a switching device in accordance with  FIG. 2A , with n=4, 5, or 6 while keeping the total transmission line  202 A length and the total Cshunt capacitance the same. Utilizing a more conservative power point before insertion loss begins to significantly decline, the 1.5 dB bandwidth (indicated by corresponding markers m1, m2, and m3) for the three circuit variations improved from 21.5 GHz (n=4, see line  302 ) to 25.4 GHz (n=5, see line  304 ) to 29.9 GHz (n=6, see line  306 ) as n increased. 
     Further, working with equations Eq. 2 and Eq. 3, the value of Cshunt can be expressed in terms of the desired Zon, f c , and number of networks n as follows: 
     
       
         
           
             
               
                 
                   Cshunt 
                   = 
                   
                     n 
                     
                       2 
                       ⁢ 
                       π 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         f 
                         c 
                       
                       * 
                       Zon 
                     
                   
                 
               
               
                 
                   [ 
                   
                     Eq 
                     . 
                     
                         
                     
                     ⁢ 
                     4 
                   
                   ] 
                 
               
             
           
         
       
     
     Therefore, the maximum Cshunt can be calculated for a set of targeted parameters. As an example, for a Zon of 50 Ohms, a cutoff frequency of 60 GHz, and n=6 for the number of tuning networks  206 , results in Cshunt=318 fF, Cshunt/n=53 fF, and L 1 =66 pH. 
     (3) Improving Isolation of OFF Paths 
     For the configuration shown in  FIG. 2A , as noted above, the higher the number n of shunt switch units  204  and corresponding inductive tuning components  203  (i.e., tuning networks  206 ), the higher the cutoff frequency. In addition, as n increases, the higher the number of LR low pass filter stages there are in the OFF path, and accordingly the isolation of the OFF path (from RFC to RF 2 ) is improved compared to conventional designs. 
     Stacked Switch Structures 
     As mentioned above, each of the shunt switch units  204  and the series switches  201 A,  201 B may be replaced by multiple series-coupled FET switches. This type of “stacked” architecture allows a circuit to tolerate higher voltages than a single FET switch. For example,  FIG. 4A  is a schematic diagram of a circuit architecture having distributed stacked shunt switches as well as distributed gate resistors (the gate resistor aspect is discussed below). In this example, each of the n shunt switch units  204  of  FIG. 2A  has been replaced by a series-coupled stack of m FET switches  402 , where m≧2. 
     For some embodiments that may not require distributed shunt switches, a lumped design with stacked shunt switches may be used. For example,  FIG. 4B  is a schematic diagram of a circuit architecture having lumped stacked shunt switches. In this example, the n shunt switch units  204  of  FIG. 2A  has been replaced by a single series-coupled stack of m FET switches  404 . 
     The series switches  201 A,  201 B shown in  FIG. 2A  may also be implemented as stacked switches. For example,  FIG. 5  is a schematic diagram of a circuit architecture  500  having stacked series switches. One or more of the series switches  201 A,  201 B shown in  FIG. 2A  would be replaced by two or more FET switches  502  configured as a series-coupled stack. 
     Symmetrical Layout 
     The switching device architecture shown in  FIG. 2A  can be advantageously combined with the stacked switch circuits shown in  FIG. 4A ,  FIG. 4B , and/or  FIG. 5  to provide a distributed stacked FET-switch based switching device that provides for approximately even electromagnetic field distribution around the transmission lines  202 A,  202 B, and provides for a better ground return. 
     For example,  FIG. 6  is a diagram of a conceptual circuit layout of an RF switching circuit  600  with distributed switches (which may be stacked switches) for both the shunt and series switch components. In the illustrated embodiment, two terminal ports  102 A,  102 B (shown series connected to respective external loads RF 1 , RF 2 ) are connectable to a common port  104  (shown series connected to an external load RFC) through corresponding transmission lines  202 A,  202 B and series switches  602 A,  602 B. The series switches  602 A,  602 B may be single FET switches as shown in  FIG. 2A , or a stack of FET switches as shown in  FIG. 5 . As in  FIG. 2A  and  FIG. 2C , the transmission lines  202 A,  202 B each include a plurality of inductive tuning components  203 , and, optionally, supplemental inductive tuning components  207  (not shown). 
     Coupled to the transmission lines  202 A,  202 B are sets of n shunt switches  604 , each of which may be configured as shown in  FIG. 2A  (distributed),  FIG. 4A  (stacked distributed), or  FIG. 4B  (lumped distributed); in each configuration, there is an internal connection to circuit ground (not shown in  FIG. 6 ). As discussed above, a configuration with n shunt switch units alone or in conjunction with control of the inductance values of the inductive tuning components  203 ,  207  gives control over the cutoff frequency of the switching circuit  600 , provides the ability to tune out the effects of Cshunt and Coff, and improves isolation of OFF paths. In addition, by using a stack of m FET switches for each of the n shunt switch units, higher voltage levels can be tolerated. 
     Importantly, in the configuration shown in  FIG. 6 , the sets of shunt switches  604  are physically placed on both sides of the transmission lines  202 A,  202 B, and the transmission lines  202 A,  202 B are arrayed on an IC layout in a substantially symmetrical manner. Such placement of the sets of shunt switches  604  provides for approximately even electromagnetic field distribution around the transmission lines  202 A,  202 B and provides for a better ground return because of the multiple connections to circuit ground; both characteristics are useful when designing absorptive switches. In addition, such physical distribution improves the thermal characteristics of the switching device  600  by spacing apart the FET switches, thus reducing the areal concentration of power-consuming circuit elements. 
     As noted above, in other configurations, only one terminal port may be used (e.g., a single-pole, single-throw switch), more than two terminal ports may be included (e.g., a 1×N switch), and more than one common port may be included (e.g., an M×N matrix switch). Accordingly, fewer or additional transmission lines may be arrayed on an IC layout in a substantially symmetrical manner as needed to accommodate fewer or additional ports, with associated sets of shunt switches  604  physically placed on both sides of the added transmission lines. 
     Gate Resistance Area Reduction 
     In general, FET switches require a gate resistor to limit the instantaneous current that is drawn when the FET is turned on, to control the switch ON and OFF times, and in general to maintain electromagnetic integrity. In conventional IC FET designs, a gate resistor is physically located in close proximity to the gate of the transistor. However, when implementing a distributed shunt switch of the type shown in  FIG. 4A , each FET shunt switch unit  204  is n times smaller than in a lumped design. Accordingly, to maintain the same low frequency characteristics, the gate resistor value for each FET shunt switch unit  204  must be n times larger than in a lumped design. Further, each FET shunt switch unit  204  may include m FET switches  402 . In such a configuration, since there are m stacked elements per shunt switch unit  204 , the total size for all of the gate resistors is m*n times bigger in area than in a lumped design. 
     To reduce the total size of the needed gate resistance, in some embodiments a FET gate resistor can be split into two sections. Referring again to  FIG. 4A , each of the FET switches  402  includes a small primary resistance R 1  (e.g., about from 1,000 to 1,000,000 ohms) that is placed in close proximity to the gate of each FET to take care of needed electromagnetic integrity. A larger secondary resistance R 2  (e.g., about from 10,000 to 10,000,000 ohms) is then placed on a common path series coupled to multiple instances of the small primary resistance R 1  (e.g., each of the n FET shunt switch units  204 ) to maintain desired low frequency characteristics, but may be physically located away from close proximity to the small primary resistances R 1  (and hence from the gate of each FET). The values for R 1  and R 2  are set such that R 1 / n +R 2 =R, where R is the total gate resistance needed for a particular circuit design. Such values may be empirically determined by experiment or simulation for frequencies of interest. Because the larger secondary resistances R 2  are shared over a number of FET switches, the total area needed for integrated circuit fabrication will be reduced in scale by the ratio of R 1  to R 2 . As should be apparent, each primary resistance R 1  and secondary resistance R 2  may comprise two or more actual resistive elements. 
     Methods 
     Another aspect of the invention includes a method for configuring a radio frequency switching device, including the steps of: 
     STEP 1: providing at least one common port; 
     STEP 2: providing at least one field effect transistor (FET) series switch, each coupled to at least one common port; 
     STEP 3: providing at least one transmission line, each coupled to a respective FET series switch, each transmission line including at least one series-coupled inductive tuning component; 
     STEP 4: providing at least one terminal port, each coupled to a respective transmission line; and 
     STEP 5: providing, for each transmission line, at least one FET shunt switch unit coupled to circuit ground and to such transmission line in a tuning network configuration. 
     A further aspect of the invention includes a method for configuring a radio frequency switching device, including the steps of: 
     STEP 1: providing at least one common port; 
     STEP 2: coupling at least one field effect transistor (FET) series switch to at least one common port; 
     STEP 3: coupling at least one transmission line to a respective FET series switch, each transmission line including at least one series-coupled inductive tuning component; 
     STEP 4: coupling at least one terminal port to a transmission line; and 
     STEP 5: coupling at least one FET shunt switch unit to circuit ground and to each such transmission line in a tuning network configuration. 
     The described method can be extended to include physically positioning pairs of the FET shunt switch units on both sides of each of the at least one transmission line; arraying the at least one transmission line on an integrated circuit layout in a substantially symmetrical manner; configuring at least one FET shunt unit as a series-coupled stack of FET switches; configuring at least one FET series switch as a series-coupled stack of FET switches; coupling at least one primary resistor to a gate of each FET in the FET shunt unit in close proximity to such gate, and providing a plurality of secondary resistors each series coupled to the primary resistors of two or more FETs but located farther away from the gate of each such FET than the primary resistors coupled to each such gate; coupling at least one secondary isolation FET series switch between a respective one of the at least one transmission line and a respective one of the at least one terminal port; and fabricating the described circuitry as an integrated circuit. 
     As should be readily apparent to one of ordinary skill in the art, various embodiments of the invention can be implemented to meet a wide variety of specifications. Thus, selection of suitable component values are a matter of design choice unless otherwise noted. The switching and passive elements may be implemented in any suitable integrated circuit (IC) technology, including but not limited to MOSFET and IGFET structures. Integrated circuit embodiments may be fabricated using any suitable substrates and processes, including but not limited to standard bulk silicon, silicon-on-insulator (SOI), silicon-on-sapphire (SOS), GaAs pHEMT, and MESFET processes. Voltage levels may be adjusted or voltage polarities reversed depending on a particular specification and/or implementing technology (e.g., NMOS, PMOS, or CMOS). Component voltage, current, and power handling capabilities may be adapted as needed, for example, by adjusting device sizes, “stacking” components to tolerate greater voltages (including as described above), and/or using multiple components in parallel to tolerate greater currents. Additional circuit components may be added to enhance the capabilities of the disclosed circuits and/or to provide additional functional without significantly altering the functionality of the disclosed circuits. 
     A number of embodiments of the invention have been described. It is to be understood that various modifications may be made without departing from the spirit and scope of the invention. For example, some of the steps described above may be order independent, and thus can be performed in an order different from that described. Various activities described with respect to the methods identified above can be executed in repetitive, serial, or parallel fashion. It is to be understood that the foregoing description is intended to illustrate and not to limit the scope of the invention, which is defined by the scope of the following claims, and that other embodiments are within the scope of the claims.