Patent Publication Number: US-8525436-B2

Title: Light-emitting diode (LED) current balance circuit

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The invention relates to a light-emitting diode (LED) driving circuit. More particularly, the invention relates to an LED current balance circuit for driving a plurality of lightbars each including a plurality of LEDs coupled in series. 
     2. Description of the Related Art 
       FIG. 1  is a diagram illustrating a conventional LED current balance circuit for a single lightbar. Referring to  FIG. 1 , a lightbar  11  includes a plurality of LEDs D 1 -Dn coupled in series, where n is a positive integer. A forward voltage Vf 1  of the lightbar  11  is a total sum of forward voltages of the LEDs D 1 -Dn. The lightbar  11  has a first terminal coupled to a lightbar voltage VBUS and a second terminal coupled to an LED current balance circuit  1 . The LED current balance circuit  1  includes a transistor Q, a resistor R and an operational amplifier OP. The resistor R detects a current through the lightbar  11  and accordingly generates a detecting result signal. The operational amplifier OP has an inverting terminal for receiving the detecting result signal (corresponding to the current through the lightbar  11 ), a non-inverting terminal for receiving a setting signal Vset (corresponding to a desired current) and an output terminal for outputting a control signal according to the difference between the current through the lightbar  11  and the desired current. 
     The transistor Q adjusts, according to the control signal outputted from the operational amplifier OP, its operating point to adjust a voltage drop across the transistor Q to further adjust the forward voltage Vf 1  to cause the current through the lightbar  11  to follow the desired current. Therefore, a brightness of the lightbar  11  (corresponding to the current through the lightbar  11 ) is kept substantially constant and equal to a desired brightness (corresponding to the desired current) regardless of a tolerance of the forward voltage Vf 1  of the lightbar  11  which is a total sum of tolerances of the forward voltages of the LEDs Dl-Dn even though they are manufactured in the same batch and the same process by the same manufacturer. 
       FIG. 2  is a diagram illustrating a conventional LED current balance circuit for a plurality of lightbars. Referring to  FIG. 2 , a plurality of lightbars  11 - 1 m are employed, and each lightbar  1 i is constructed such as the lightbar  11  shown in  FIG. 1 , where m is a positive integer and i is an integer from 1 to m. Each lightbar  1 i has a first terminal coupled to a lightbar voltage VBUS and a second terminal coupled to a current balance unit (not shown) constructed such as the LED current balance circuit  1  shown in  FIG. 1 . The LED current balance circuit  2  includes a direct-current to direct-current (DC/DC) converter  21  and an LED controller  22 . The DC/DC converter  21  converts a DC input voltage VIN to the lightbar voltage VBUS. The LED controller  22  includes the m current balance units coupled to the lightbars  11 - 1 m through a plurality of channel terminals CH 1 -CHm. Accordingly, a current through each lightbar  1 i will follow a desired current such as the desired current (corresponding to the setting signal Vset) shown in  FIG. 1 . In other words, the currents through the lightbars  11 - 1 m are kept substantially constant and equal to each other to achieve current balance. However, as the number of the lightbars  11 - 1 m increases, the number of the m current balance units increases, resulting in a cost and circuit size increase, and the effect of current balance becomes worse due to the increase in the number of the current balance units having their respective tolerance. 
     Recently, many specific-purpose integrated circuits (ICs) for the LED controller  22  have been developed. The specific-purpose LED controller IC integrates a fixed number of the current balance units and other functional units. For instance, a functional unit (not shown) in the LED controller  22  outputs a feedback signal through a feedback terminal FB to control the DC/DC converter  21  to adjust the lightbar voltage VBUS to provide feedback control to optimize the lightbar voltage VBUS applied to the lightbars  11 - 1 m. Although the specific-purpose LED controller IC has the advantage of more accurate control and smaller circuit size, it has the disadvantage of higher cost, lower reliability and limited current and power ratings (typically less than 60mA). In a high-voltage and large-current LED lightbar application, there is a need for the specific-purpose LED controller IC to employ an external control manner by using some external components such as m transistor and m resistor for m lightbars so that cost and circuit size also increase as the number of the lightbars increases. 
     SUMMARY OF THE INVENTION 
     Accordingly, an LED current balance circuit is provided for driving a plurality of lightbars each including a plurality of LEDs coupled in series and for balancing currents through the lightbars by a simpler circuit architecture. 
     According to an aspect of the invention, an LED current balance circuit is provided. The LED current balance circuit drives a plurality of lightbars. Each lightbar includes a plurality of LEDs coupled in series, and each lightbar has a first terminal coupled to a lightbar voltage and a second terminal. The LED current balance circuit includes a current mirror, a reference current generator and a voltage compensation circuit. The current mirror balances currents through the lightbars by generating a plurality of sink currents according to a reference current and by causing each sink current to sink a current from the second terminal of a corresponding lightbar while the current mirror is enabled. The current mirror causes the currents through the lightbars to be zero while the current mirror is disabled. The reference current generator is coupled to the current mirror and is supplied power from a first supply voltage. The reference current generator provides the reference current and a second supply voltage. The reference current varies according to the first supply voltage while the first supply voltage is less than a constant-current threshold value to implement an analog dimming by enabling the current mirror and employing the first supply voltage with variable voltage as a dimming signal. The reference current is constant while the first supply voltage is greater than the constant-current threshold value to implement a digital dimming by employing a pulse-width modulation (PWM) signal with a variable pulse width as the dimming signal to alternatively enable and disable the reference current generator or the current mirror. The voltage compensation circuit is coupled to the second terminals of the lightbars and is supplied power from the second supply voltage. The voltage compensation circuit adjusts the lightbar voltage down while a voltage at the second terminal of one of the lightbars is greater than a compensation threshold value, and adjusts the lightbar voltage up while voltages at the second terminals of the lightbars are less than the compensation threshold value under the condition of the turn-on of all lightbars. 
     In one embodiment, the reference current generator includes a first bipolar junction transistor (BJT), an adjustable shunt regulator, a first resistor and a second resistor. The adjustable shunt regulator has a cathode terminal, an anode terminal and a reference terminal. A collector terminal of the first BJT is coupled to the first supply voltage and a first terminal of the first resistor. A base terminal of the first BJT is coupled to a second terminal of the first resistor and the cathode terminal of the adjustable shunt regulator. An emitter terminal of the first BJT is coupled to the reference terminal of the adjustable shunt regulator and a first terminal of the second resistor. The anode terminal of the adjustable shunt regulator is coupled to ground. A first terminal of the second resistor provides the second supply voltage. A second terminal of the second resistor provides the reference current. 
     In one embodiment, the voltage compensation circuit includes a plurality of first diodes, a second BJT, a third resistor, a fourth resistor, a fifth resistor, a sixth resistor, a seventh resistor, an eighth resistor and a first capacitor. A cathode terminal of each first diode is coupled to the second terminal of a corresponding lightbar. An anode terminal of each first diode is coupled to a first terminal of the third resistor and a first terminal of the fourth resistor. A second terminal of the third resistor is coupled to the second supply voltage and a first terminal of the fifth resistor. A second terminal of the fourth resistor and a second terminal of the fifth resistor are coupled to a base terminal and a collector terminal of the second BJT respectively. A first terminal and a second terminal of the sixth resistor are coupled to an emitter terminal of the second BJT and ground respectively. A first terminal of the seventh resistor is coupled to the collector terminal of the second BJT and a first terminal of the first capacitor. A second terminal of the seventh resistor is coupled to a first terminal of the eighth resistor. A second terminal of the eighth resistor is coupled to a second terminal of the first capacitor and ground. The first terminal of the eighth resistor provides a compensation signal for adjusting the lightbar voltage. 
     In the invention, the reference current generator provides the reference current robust against disturbance in the (first) supply voltage applied to the reference current generator. The current mirror balances the currents through the lightbars according to the reference current. The voltage compensation circuit detects the voltages across the lightbars to compensate the lightbars having various forward voltages to ensure the turn-on of all lightbars and effectively balance the currents through the lightbars. Therefore, the invention employs a simpler circuit architecture and does not need a specific-purpose LED controller, to be cheaper and more competitive. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The foregoing and other features of the disclosure will be apparent and easily understood from a further reading of the specification and claims and by reference to the accompanying drawings in which: 
         FIG. 1  is a diagram illustrating a conventional LED current balance circuit for a single lightbar; 
         FIG. 2  is a diagram illustrating a conventional LED current balance circuit for a plurality of lightbars; 
         FIGS. 3 and 4  are a block diagram and a circuit diagram, respectively, illustrating an LED current balance circuit according to a preferred embodiment of the invention; 
         FIG. 5  is a diagram illustrating the symbol and functional block diagram of the adjustable shunt regulator shown in  FIG. 4 ; 
         FIGS. 6A and 6B  are waveform diagrams illustrating simulation results for the LED current balance circuit shown in  FIG. 4 ; and 
         FIG. 7  is a waveform diagram illustrating experimental results for the LED current balance circuit shown in  FIG. 4 . 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Those skilled in the art should understand that a NPN bipolar junction transistor (BJT) has a first terminal (i.e. a collector terminal), a second terminal (i.e. an emitter terminal) and a control terminal (i.e. a base terminal); a N-channel field-effect transistor (FET) has a first terminal (i.e. a drain terminal), a second terminal (i.e. a source terminal) and a control terminal (i.e. a gate terminal); a resistor and a capacitor each has a first terminal and a second terminal; and, a diode, an LED and a Zener diode each has an anode terminal and a cathode terminal, which are not described again in the text which follows. 
       FIGS. 3 and 4  are a block diagram and a circuit diagram, respectively, illustrating an LED current balance circuit according to a preferred embodiment of the invention. Referring to  FIGS. 3 and 4 , an LED current balance circuit  3  drives a plurality of lightbars  11 - 1 m, and each lightbar  1 i includes a plurality of LEDs D 1 -Dn coupled in series, where m and n are positive integers, and i is an integer from 1 to m, which are not described again in the text which follows. Each lightbar  1 i has a first terminal coupled to a lightbar voltage VBUS and a second terminal. Those skilled in the art should understand that the lightbars  11 - 1 m will work well if each lightbar  1 i satisfies that an anode terminal of the LED D 1  is coupled to the first terminal of the lightbar  1 i , a cathode terminal of the LED Dk is coupled to an anode terminal of the LED D(k+1), and a cathode terminal of the LED Dn is coupled to the second terminal of the lightbar  1 i , where k is an integer from 1 to (n−1). The lightbars  11 - 1 m may be employed as a direct-light-type or edge-light-type backlight of a liquid crystal display (LCD). 
     The LED current balance circuit  3  includes a DC/DC converter  31 , a reference current generator  32 , a current mirror  33 , a voltage compensation circuit  34 , an overvoltage detector  35  and a dimming circuit  36 . 
     The DC/DC converter  31  such as a bulk or boost converter converts a DC input voltage VIN such as 5V, 12V or 24V provided by a general-purpose power supply (not shown) to the lightbar voltage VBUS sufficient to drive the lightbars  11 - 1 m. The DC/DC converter  31  further receives a power on/off signal Von-off, a fault signal Vfault and a compensation signal Vcomp. The power on/off signal Von-off at high level, for example, enables the DC/DC converter  31  so that the DC/DC converter  31  provides the lightbar voltage VBUS to supply power to the lightbars  11 -m and a first supply voltage VCC to supply power to internal components of the LED current balance circuit  3 . Conversely, the power on/off signal Von-off at low level disables the DC/DC converter  31  so that the DC/DC converter  31  does not supply power anymore. 
     The reference current generator  32  is coupled to the current mirror  33  and supplied power from the first supply voltage VCC. In the embodiment, the reference current generator  32  includes a first BJT Q 1 , an adjustable shunt regulator TL 1 , a first resistor R 1  and a second resistor R 2 . The adjustable shunt regulator TL 1  is a TL431 IC manufactured by Texas Instruments Inc.  FIG. 5  is a diagram illustrating the symbol and functional block diagram of the adjustable shunt regulator TL 1  shown in  FIG. 4 . Referring to  FIG. 5 , the adjustable shunt regulator TL 1  has a cathode terminal, an anode terminal and a reference terminal. The adjustable shunt regulator TL 1  includes a constant voltage source, a transistor Q and an operational amplifier OP. The constant voltage source provides a reference voltage VREF (typically 2.5V for TL431). The operational amplifier OP has an inverting terminal coupled to the reference voltage VREF, a non-inverting terminal coupled to the reference terminal and an output terminal coupled to a base terminal of the transistor Q. A collector terminal and an emitter terminal of the transistor Q are coupled to the cathode terminal and the anode terminal respectively. A current through the transistor Q is a stable non-saturation current while a voltage at the reference terminal is very close to the reference voltage VREF, and the current through the transistor Q varies within a range of 1 A to 100 mA as the voltage at the reference terminal slightly varies relative to the reference voltage VREF. 
     Referring back to  FIGS. 3 and 4 , in the reference current generator  32 , a collector terminal of the first BJT Q 1  is coupled to the first supply voltage VCC and a first terminal of the first resistor R 1 . A base terminal of the first BJT Q 1  is coupled to a second terminal of the first resistor R 1  and the cathode terminal of the adjustable shunt regulator TL 1 . An emitter terminal of the first BJT Q 1  is coupled to the reference terminal of the adjustable shunt regulator TL 1  and a first terminal of the second resistor R 2 . The anode terminal of the adjustable shunt regulator TL 1  is coupled to ground. A first terminal of the second resistor R 2  provides a second supply voltage VEE for supplying power to the voltage compensation circuit  34 . A second terminal of the second resistor R 2  provides a reference current Iref. 
     While the first supply voltage VCC is greater than a constant-current threshold value, the first BJT Q 1  is conducted, and the adjustable shunt regulator TL 1  is normally operated, so that a voltage at the reference terminal of the adjustable shunt regulator TL 1  (i.e. the second supply voltage VEE) is substantially a constant voltage. If a resistance of the second resistor R 2  is determined, the reference current Iref provided from the reference current generator  32  will be determined and substantially constant, and a voltage at the second terminal of the second resistor R 2  is determined and substantially constant. The current mirror  33  is operated more stably while being biased by the constant current (Iref) and the constant voltage provided from the reference current generator  32 . In the embodiment, a collector-emitter voltage of the first BJT Q 1  while being conducted is about 1V, the voltage at the reference terminal of the adjustable shunt regulator TL 1  is about 2.5V, and the constant-current threshold value is therefore about 3.5V. In addition, the first supply voltage VCC, of course, has an upper limit which is mainly determined by maximum current and power ratings of the first BJT Q 1 . 
     The current mirror  33  is coupled to the reference current generator  32  and the second terminals of the lightbar  11 - 1 m. While the current mirror  33  is enabled, the current mirror  33  balances currents through the lightbars  11 - 1 m by generating a plurality of sink currents  11 - 1 m according to the reference current Iref and by causing each sink current  1 i to sink a current from the second terminal of a corresponding lightbar  1 i. While the current mirror  33  is disabled, the current mirror  33  causes the currents through the lightbars  11 - 1 m to be zero. 
     In the embodiment, the current mirror  33  includes a plurality of first transistors Q 11 -Q 1 m matched to each other and a second transistor Q 22 . The first transistors Q 11 -Q 1 m and the second transistor Q 22  employed here are, but are not limited to, NPN BJTs. The transistors Q 11 -Q 1 m and Q 22  may be, for example, N-channel FETs. The first terminal of each first transistor Q 11  is coupled to the second terminal of a corresponding lightbar  1 i for sinking a corresponding sink current  1 i from the corresponding lightbar  1 i . The first terminal and the control terminal of the second transistor Q 22  are coupled to each other so that the second transistor Q 22  forms a diode-connected transistor. The first terminal of the second transistor Q 22  is further coupled to the second terminal of the second resistor R 2  of the reference current generator  32  for receiving the reference current Iref. The second terminals of the first transistors Q 11 -Q 1 m and the second transistor Q 22  are coupled to ground. The control terminals of the first transistors Q 11 -Q 1 m and the second transistor Q 22  are coupled to each other. While the current mirror  33  is enabled, the sink currents I 1 -Im are substantially equal to each other due to the matched first transistors Q 11 -Q 1 m to cause the currents through the lightbars  11 - 1 m to be substantially equal to each other so that the lightbars  11 - 1 m can provide stable and uniform brightness. 
     In the embodiment, the second terminal of each first transistor Q 1 i is coupled to ground through a corresponding resistor R 1 i, and the second terminal of the second transistor Q 22  is coupled to ground through a corresponding resistor R 22 . The resistors R 11 -Rlm can reduce the influence of the unmatched first transistors Q 11 -Q 1 m on the sink currents I 1 -Im. While the control terminal of the second transistor Q 22  is coupled to ground, the first transistors Q 11 -Q 1 m and the second transistor Q 22  are cut off to achieve the disablement of the current mirror  33 . While the control terminal of the second transistor Q 22  is not coupled to ground, the first transistors Q 11 -Q 1 m and the second transistor Q 22  are operated normally to achieve the enablement of the current mirror  33 . 
     The voltage compensation circuit  34  includes a plurality of first diodes D 11 -D 1 m, a second BJT Q 2 , a third resistor R 3 , a fourth resistor R 4 , a fifth resistor R 5 , a sixth resistor R 6 , a seventh resistor R 7 , an eighth resistor R 8  and a first capacitor C 1 . A cathode terminal of each first diode D 1 i is coupled to the second terminal of a corresponding lightbar  1 i. An anode terminal of each first diode D 1 i is coupled to a first terminal of the third resistor R 3  and a first terminal of the fourth resistor R 4 . A second terminal of the third resistor R 3  is coupled to the second supply voltage VEE and a first terminal of the fifth resistor R 5 . A second terminal of the fourth resistor R 4  and a second terminal of the fifth resistor R 5  are coupled to a base terminal and a collector terminal of the second BJT Q 2  respectively. A first terminal and a second terminal of the sixth resistor R 6  are coupled to an emitter terminal of the second BJT Q 2  and ground respectively. A first terminal of the seventh resistor R 7  is coupled to the collector terminal of the second BJT Q 2  and a first terminal of the first capacitor C 1 . A second terminal of the seventh resistor R 7  is coupled to a first terminal of the eighth resistor R 8 . A second terminal of the eighth resistor R 8  is coupled to a second terminal of the first capacitor C 1  and ground. The first terminal of the eighth resistor R 8  provides the compensation signal Vcomp for adjusting the lightbar voltage VBUS. 
     It is assumed that each lightbar  1 i includes 13 LEDs and that a voltage at the second terminal of each lightbar  1 i (or a voltage across the first transistor Q 1 i and the resistor R 1 i of the current mirror  33 ) is 1V. Ideally, a forward voltage of each LED is 3.3V so that a forward voltage Vfi of the lightbar  1 i is 42.9V (=3.3V×13) and so that the lightbar voltage VBUS is 43.9V. However, in fact, the forward voltages Vf 1 -Vfm of the lightbars  11 - 1 m are different. Some (such as the lightbar  11 ) are greater than 42.9V, and the others (such as the lightbar  12 ) are slightly less than 42.9V. To ensure the turn-on of all lightbars  11 - 1 m, the lightbar voltage VBUS should be slightly greater than 43.9V to turn on the lightbar  11  (which is a representative of the lightbars whose forward voltages are greater than 42.9V), but it results that the lightbar  12  (which is a representative of the lightbars whose forward voltages are less than 42.9V) has a higher voltage at its second terminal. Hence, the total voltage drop across the first transistor Q 11  and the resistor R 11  becomes larger to dissipate more power and reduce useful life. The invention employs the voltage compensation circuit  34  to adjust the lightbar voltage VBUS to reduce the power dissipation of the current mirror  33  under the condition of the turn-on of all lightbars  11 - 1 m. 
     It is assumed that a forward voltage of each first diode D 1 i is 0.7V. Ideally, the voltage at the second terminal of each lightbar  1 i is 1V, and, then, a voltage at the anode terminal of each first diode D 1 i is 1.7V. While a voltage at the second terminal of one of the lightbars  11 - 1 m (or a total voltage drop across one first transistor and a corresponding resistor) is greater than 1V to bring more power dissipation in the current mirror  33 , the voltage at the anode terminal of the first diode D 1 i is greater than 1.7V (defined as a compensation threshold value). Accordingly, the second BJT Q 2  is designed to be conducted to cause a voltage across the capacitor C 1  to fall so that the voltage across the capacitor C 1  is divided by the resistors R 7  and R 8  to generate the compensation signal Vcomp. Conversely, while the voltages at the second terminals of the lightbars  11 - 1 m are all less than 1V, the voltage at the anode terminal of the first diode D 1 i is less than 1.7V, and the second BJT Q 2  is cut off so that the second supply voltage VEE is divided by the resistors R 5 , R 7  and R 8  to generate the compensation signal Vcomp. According to the compensation signal Vcomp, under the condition of the turn-on of all lightbars  11 - 1 m, the DC/DC converter  31  decreases or adjusts the lightbar voltage VBUS down while the voltages at the second terminals of the lightbars  11 - 1 m are all less than the compensation threshold value, and increases or adjusts the lightbar voltage VBUS up while the voltage at the second terminal of one of the lightbars  11 - 1 m is greater than the compensation threshold value. In addition, the compensation threshold value can be expressed as (VEE-Vbe 2 )×[R 4 +(1+β)×R 6 ]/[R 3 +R 4 +(1+β)×R 6 ], where Vbe 2  is a base-emitter voltage of the second BJT Q 2  and  13  is a common-emitter current gain of the second BJT Q 2 . 
     The overvoltage detector  35  is coupled to the second terminals of the lightbars  11 - 1 m. The overvoltage detector  35  provides the fault signal Vault while the overvoltage detector  35  detects if a voltage at the second terminal of one of the lightbars  11 - 1 m is greater than an overvoltage threshold value, and the fault signal Vault causes the lightbar voltage VBUS to be zero. 
     In the embodiment, the overvoltage detector  35  includes a plurality of second diodes D 21 -D 2 m, a Zener diode ZD 1 , a ninth resistor R 9 , a tenth resistor R 10  and a second capacitor C 2 . An anode terminal of each second diode D 2 i is coupled to the second terminal of a corresponding lightbar  1 i. A cathode terminal of each second diode D 2 i is coupled to a cathode terminal of the Zener diode ZD 1 . An anode terminal of the Zener diode ZD 1  is coupled to a first terminal of the ninth resistor R 9 . A second terminal of the ninth resistor R 9  is coupled to a first terminal of the tenth resistor R 10  and a first terminal of the second capacitor C 2 . A second terminal of the tenth resistor R 10  is coupled to a second terminal of the second capacitor C 2  and ground. The first terminal of the tenth resistor R 10  provides the fault signal Vfault while the overvoltage detector  35  detects if a voltage at the second terminal of one of the lightbars  11 - 1 m is greater than the overvoltage threshold value. While a voltage at the second terminal of one lightbar (such as the lightbar  11 ) is greater than the overvoltage threshold value, the Zener diode ZD 1  is operated in a reverse breakdown region, and a voltage which is the voltage at the second terminal of the lightbar  11  minus the sum of the forward voltage of the second diode D 21  and the breakdown voltage of the Zener diode ZD 1  is dropped across the ninth resistor R 9  and the tenth resistor R 10 , so that the voltage dropped across the tenth resistor R 10  is designed to be a high level to represent that the fault signal Vfault is outputted. Converserly, while the voltages at the second terminals of the lightbars  11 - 1 m are all less than the overvoltage threshold value, the Zener diode ZD 1  is operated in a reverse region but not in a reverse breakdown region, and there is no voltage dropped across the ninth resistor R 9  and the tenth resistor R 10 , so that the voltage dropped across the tenth resistor R 10  is zero or at low level to represent that the fault signal Vfault is not outputted. In addition, the overvoltage threshold value can be determined by using the Zener diode ZD 1  with a specific breakdown voltage. 
     While the first supply voltage VCC is greater than the constant-current threshold value, the reference current Iref provided from the reference current generator  32  is constant so that the sink currents I 1 -Im are constant to cause the currents through the lightbars  11 - 1 m to be constant while the current mirror  33  is enabled. That is, the lightbars  11 - 1 m provide constant brightness. The LED current balance circuit  3  must employ a digital dimming to implement the brightness adjustment function of the lightbars  11 - 1 m. The digital dimming is implemented by alternatively enabling and disabling the current mirror  33  (or the reference current generator  32 ) to alternatively turn on and turn off the lightbars  11 - 1 m in turn. By using the persistence of vision, a human eye may only perceive an average brightness based on the ratio of the turn-on period and the turn-off periods period of the lightbars  11 - 1 m (corresponding to the ratio of the enablement period and the disablement periods period of the current mirror  33  or the reference current generator  32 ). 
     In the embodiment, the current mirror  33  is alternatively enabled and disabled through the dimming circuit  36 . The dimming circuit  36  is coupled to the reference current generator  32  and the current mirror  33 . The dimming circuit  36  receives a pulse-width modulation (PWM) signal Vpwm through a dimming terminal DIM of the dimming circuit  36  and alternatively enables and disables the current mirror  33  according to the PWM signal Vpwm. The ratio of the enablement period and the disablement period of the current mirror  33  can be determined by adjusting a pulse width (or a duty cycle) of the PWM signal Vpwm. Accordingly, the digital dimming is implemented by employing the PWM signal Vpwm with a variable pulse width as a dimming signal. 
     In the embodiment, the dimming circuit  36  includes transistor switches Q 3 -Q 6  and resistors R 31 -R 34 . While the PWM signal Vpwm is at a low level, the transistor switch Q 5  is turned off and the transistor switch Q 6  is turned on so that the base terminals of the transistors Q 11 -Q 1 m and Q 22  of the current mirror  33  are coupled to ground, the transistors Q 11 -Q 1 m and Q 22  are cut off, the sink currents I 11 -I 1 m are not generated, and the current mirror  33  is disabled. While the PWM signal Vpwm is at a high level, the transistor switch Q 5  is turned on, and the transistor switch Q 6  is turned off, so that the dimming circuit  36  does not influence the operation of the current mirror  33  and so that the current mirror  33  is enabled. Moreover, while the power on/off signal Von-off is at a low level, the transistor switch Q 3  is turned off, and the transistor switch Q 4  is turned on, so that the base terminal of the first BJT Q 1  of the reference current generator  32  is coupled to the ground, the first BJT Q 1  is cut off, the cathode terminal and the anode terminal of the adjustable shunt regulator TL 1  are coupled to ground, the reference current Iref and the second supply voltage VEE are zero, and the reference current generator  32  is disabled. While the power on/off signal Von-off is at a high level, the transistor switch Q 3  is conducted, and the transistor switch Q 4  is cut off, so that the dimming circuit  36  does not influence the operation of the reference current generator  32  and so that the reference current generator  32  is enabled. 
       FIGS. 6A and 6B  are waveform diagrams illustrating simulation results for the LED current balance circuit shown in  FIG. 4 . It is simulated under the condition that the LED current balance circuit  3  drives 6 lightbars  11 - 16 , the currents  11 - 16  through the lightbars  11 - 16  each is 20 mA, and the duty cycle of the PWM signal Vpwm is 50%. First, referring to  FIG. 6A , it is simulated under the condition that the first supply voltage VCC is constant and equal to 5V. It is observed that the sink currents I 1 -I 6  (correspond to the currents through the lightbars  11 - 16 ) are kept substantially constant and equal to 20 mA to achieve current balance while the current mirror  33  is enabled. Accordingly, the lightbars  11 - 16  provide stable and uniform brightness. Next, referring to  FIG. 6B , it is simulated under the condition that the first supply voltage VCC is 5V with a disturbance within a range of 4V to 9V. It is observed that the sink currents I 1 -I 6  are kept substantially constant and equal to 20 mA so that the LED current balance circuit  3  are robust against the disturbance in the first supply voltage VCC. 
       FIG. 7  is a waveform diagram illustrating experimental results for the LED current balance circuit shown in  FIG. 4 . It is experimentally measured under the condition that the LED current balance circuit  3  drives  6  lightbars  11 - 16  and that the currents I 1 -I 6  through the lightbars  11 - 16  each is 20 mA. Referring to  FIG. 7 , it is a measured waveform diagram illustrating a current through one lighbar under the condition that the duty cycles of the PWM signal Vpwm are 1%, 25%, 50% and 95%, respectively. It is observed that while the duty cycle (or pulse width) of the PWM signal Vpwm is changed, the enablement period of the current mirror  33  is changed according to the pulse width of the PWM signal Vpwm. The current through the lighbar, while the current mirror  33  is enabled, is kept substantially constant and equal to 20 mA so that the LED current balance circuit  3  provides a good dimming linearity. In addition, the measured currents I 1 -I 6  through the lightbars  11 - 16  are 19.8 mA, 19.8 mA, 19.9 mA, 19.9 mA, 20.1 mA and 20.0 mA, respectively, so that the LED current balance circuit  3  provides a good current regulation of about 1.5%. 
     The above-mentioned dimming for the LED current balance circuit  3  is the digital dimming (called PWM dimming). However, the LED current balance circuit  3  can be changed to employ analog dimming. The analog dimming is implemented by enabling the current mirror  33  and employing the first supply voltage VCC with variable voltage less than the constant-current threshold value as a dimming signal. While the first supply voltage VCC is less than the constant-current threshold value, the first BJT Q 1  is cut off, and the reference current Iref varies according to the first supply voltage VCC to implement the analog dimming by employing the variable first supply voltage VCC as the dimming signal. It is noted that while employing the analog dimming, the dimming circuit  36  used in the digital dimming must be inactive by setting the PWM signal Vpwm received from the dimming terminal DIM always at high level. Alternately, the dimming circuit  36  used in the digital dimming must be removed, so that the current mirror  33  is always enabled while the LED current balance circuit  3  is operated normally. 
     Furthermore, the LED current balance circuit  3  can be changed to employ a mixing dimming combined by the analog and digital dimmings to achieve higher brightness contrast. The mixing dimming is implemented by the first supply voltage VCC being further coupled to the dimming terminal DIM and the PWM signal Vpwm being received through the dimming terminal DIM. While the duty cycle of the PWM signal Vpwm (such as 50%) is greater than a duty-cycle threshold value (such as 20%), the PWM signal Vpwm at an enablement period (such as at a high level) is a constant voltage and greater than the constant-current threshold value, the LED current balance circuit  3  behaves like the digital dimming. While the duty cycle of the PWM signal Vpwm (such as 10%) is less than the duty-cycle threshold value (such as 20%), the PWM signal Vpwm at the enablement period is a variable voltage and less than the constant-current threshold value, and the variable voltage decreases as the duty cycle of the PWM signal Vpwm. The LED current balance circuit  3  behaves like the digital dimming as well as the analog dimming to achieve higher brightness contrast under the condition lower brightness. 
     In summary, in the invention, the reference current generator provides the reference current robust against disturbance in the (first) supply voltage applied to the reference current generator. The current mirror generates, according to the reference current, the sink currents to bias the lightbars and employs a structure to reduce the influence of the unmatched first transistors on the sink currents to stabilize and clamp the currents through the lightbar. The voltage compensation circuit detects the voltages across the lightbars to compensate the lightbars having various forward voltages to ensure the turn-on of all lightbars and to effectively balance the currents through the lightbars. Therefore, the invention employs a simpler circuit architecture and does not need a specific-purpose LED controller, to be cheaper and more competitive. 
     It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the invention cover modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents.