Patent Publication Number: US-8542774-B2

Title: Receiver and method

Description:
FIELD OF THE INVENTION 
     The present invention relates to receivers and methods for detecting and recovering data from Orthogonal Frequency Division Multiplexed (OFDM) symbols that have been transmitted via a channel. 
     BACKGROUND OF THE INVENTION 
     There are many examples of communications systems in which data is communicated using Orthogonal Frequency Division Multiplexing (OFDM). Systems which have been arranged to operate in accordance with Digital Video Broadcasting (DVB) standards for example, utilise OFDM. OFDM can be generally described as providing K narrow band sub-carriers (where K is an integer) which are modulated in parallel, each sub-carrier communicating a modulated data symbol such as Quadrature Amplitude Modulated (QAM) symbol or Quadrature Phase-shift Keying (QPSK) symbol. The modulation of the sub-carriers is formed in the frequency domain and transformed into the time domain for transmission. The sub-carriers are modulated in parallel contemporaneously, so that in combination the modulated carriers form an OFDM symbol. The OFDM symbol therefore comprises a plurality of sub-carriers each of which has been modulated contemporaneously with different modulation symbols. 
     In order to allow data to be recovered from the OFDM symbols in the presence of multi-path which causes the same OFDM symbols to be received via echo paths and/or simulcast in which the same OFDM symbols are received from more than one transmitter, it is conventional to include a time domain guard interval between successive OFDM symbols. The guard interval is formed by repeating samples in the time domain from a ‘useful’ part of the OFDM symbols. The useful part of the OFDM symbols correspond to the samples in the time domain which are formed when the modulated sub-carriers are transformed into the time domain from the frequency domain. As a result of the guard interval, all of the samples from the useful part of the OFDM symbols can be received by a receiver provided that the multi-path or the simulcast delay between versions of the same OFDM symbols does not exceed the guard interval. 
     However, detecting and recovering data from the useful part of the OFDM symbols at the receiver can nevertheless present a technical problem. 
     SUMMARY OF THE INVENTION 
     According to the present invention there is provided a receiver for detecting and recovering data from Orthogonal Frequency Division Multiplexed (OFDM) symbols, the OFDM symbols including pilot data arranged in accordance with a pilot pattern and a guard interval. The guard interval is formed by copying samples from a useful part of the OFDM symbol in the time domain and a length of the guard interval for each OFDM symbol corresponds to one of a plurality of predetermined lengths. The receiver comprises a demodulator operable to detect a signal representing the OFDM symbols and to generate a sampled version of the OFDM symbols in the time domain. The receiver also includes a symbol synchronisation unit, comprising a plurality of correlators into which each sampled OFDM symbol is concurrently input, and a correlation detection processor. Each correlator is operable to auto correlate each sampled OFDM symbol between a length of samples corresponding to one of the plurality of predetermined lengths, the correlation detection processor being operable to determine a time domain start point of each OFDM symbols based on a point at which one of the correlators from the plurality of correlators detects a correlation. The receiver further comprises a frequency transform processor operable to receive the sampled version of the OFDM symbols and to perform a frequency transform on the OFDM symbol to form a frequency domain version of the OFDM symbols starting at the time domain start point determined by the correlation detect processor. The receiver also includes a coarse frequency offset detector including a pilot data filter and a coarse frequency offset detection processor. The pilot data filter includes taps corresponding to a value and spacing of the pilot pattern of the OFDM symbols and arranged to receive as an input the frequency domain OFDM symbols from the frequency transform processor. The coarse frequency detection processor is arranged to detect any coarse frequency offsets from an output of the pilot data filter. OFDM data received at a receiver is typically formed in the frequency domain and transformed into the time domain for transmission. At a receiver, in order to recover the data transmitted on each OFDM symbol, each OFDM symbol is transformed into the frequency domain typically by performing a fast Fourier transform. In order to do this accurately, a section of the OFDM symbol in the time domain must be identified which will yield an optimum amount of useful energy in the time domain on which to perform a frequency domain transform. This can be achieved by identifying the guard interval section of each OFDM symbol which occur between the OFDM symbols in the time domain. Such an identification can be achieved by correlating the received OFDM symbol with itself. However, OFDM schemes can employ guard intervals of different lengths. Conventionally therefore, before determining the section of the time domain OFDM symbol mentioned above, the guard interval length of the received OFDM symbols must be known in order to set a sample length parameter for the correlation process. Once this is known, the length of the guard interval can then be used to define a length of samples which is used to correlate the received time domain OFDM symbols to identify the position of the guard interval. The position of the OFDM symbol boundaries in the time domain can then be used to provide a symbol synchronisation timing signal which can be used to control the application of the frequency transform to the received OFDM symbols. 
     Furthermore, once the OFDM symbol has been transformed into the frequency domain, it is desirable to identify any frequency offsets on the recovered sub-carriers which are greater than the width of a sub-carrier (i.e. inter-sub-carrier offsets) in order that data can be extracted correctly from each sub-carrier, which will improve the integrity of the receiver. 
     A receiver arranged in accordance with the present invention provides an advantage in that a number of correlators, each arranged to detect correlations according to a different guard interval length, are run concurrently. Accordingly, there is no requirement to separately identify the guard interval length prior to determine the symbol boundaries which will speed up and improve an accuracy of the process of identifying the OFDM symbol boundaries. Furthermore, a coarse frequency offset detector is provided which determines a coarse frequency offset of the frequency transformed symbols by identifying a pilot sub-carrier pattern in the received symbols. This provides an advantage because a conventional feature of the received symbols (i.e. the pilot sub-carriers arranged in a pilot pattern) is used to determine the coarse frequency offset. This mitigates the requirement to include any further signalling in the OFDM symbols from which to derive the coarse frequency offset and also efficiently re-uses the pilot data which will be typically decoded in any case in order to provide a channel estimate at the receiver. 
     In one embodiment in which the sequence of OFDM symbols forms a sequence of frames and each frame is formed from one or a plurality of preamble OFDM symbols followed by a plurality of data payload bearing OFDM symbols, the receiver comprises a frame detector for detecting a beginning point of each frame by detecting the presence of pilot data carriers carrying preamble pilot data in the preamble OFDM symbols of the frame. 
     In another embodiment in which each preamble OFDM symbol comprises a plurality of header sub-carriers carrying header data, the frame detector is operable to identify a beginning point of each frame by extracting data from the sub-carriers of the received OFDM preamble symbols and identifying extracted data corresponding to the header data of the first OFDM preamble symbols. 
     Various further aspects and features of the invention are defined in the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Embodiments of the present invention will now be described by way of example only with reference to the accompanying drawings where like parts are provided with corresponding reference numerals and in which: 
         FIG. 1  provides a schematic diagram of a conventional OFDM communication network; 
         FIG. 2  provides a schematic diagram of an OFDM encoder as used in the DVB-C2 system; 
         FIG. 3  provides a schematic diagram of an OFDM DVB-C2 frame structure; 
         FIG. 4  provides a schematic diagram showing the construction of an OFDM DVB-C2 header; 
         FIG. 5  provides a schematic diagram of an L1 forward error correction and modulation unit; 
         FIG. 6  provides a schematic diagram showing functional blocks or stages of an OFDM receiver; 
         FIG. 7  provides a schematic diagram showing a symbol synchronisation unit arranged in accordance with an embodiment of the present invention; 
         FIG. 8  provides a schematic diagram indicating idealised traces of signals that flow at particular points in the symbol synchronisation detection unit shown in  FIG. 7 ; 
         FIGS. 9   a  and  9   b  provide an illustration of graphs indicating a result of a first and second stage of correlation undertaken by correlation units in the symbol synchronisation detection unit shown in  FIG. 7 ; 
         FIG. 10  provides an illustration of a technique of detecting a coarse frequency offset of frequency transformed OFDM symbols; 
         FIG. 11  provides a schematic diagram of a coarse frequency offset detector arranged to implement the technique illustrated in  FIG. 10 ; 
         FIG. 12  provides an illustration of a graph showing an input of a pulse detector of the coarse frequency offset detector shown in  FIG. 11 ; 
         FIG. 13  provides an illustration of a graph of an input of the pulse detector indicating a frame detection; 
         FIGS. 14   a  and  14   b  provide schematic diagrams illustrating a first and second implementation of a frame synchronisation detector; 
         FIG. 15  provides a diagram illustrating a graph providing an indication of the input of a pulse detector using the frame synchronisation technique implemented in the frame synchronisation detectors shown in  FIGS. 14   a  and  14   b;    
         FIGS. 16   a  and  16   b  provide schematic diagrams illustrating a third and fourth implementation of a frame synchronisation detector; 
         FIG. 17  provides a diagram illustrating a graph providing an indication of the input of a pulse detector using the frame synchronisation technique implemented in the frame synchronisation detectors shown in  FIGS. 16   a  and  16   b , and 
         FIG. 18  is a flow diagram representing a process undertaken in accordance with an embodiment of the present invention. 
     
    
    
     DESCRIPTION OF EXAMPLE EMBODIMENTS 
     Example embodiments of the present invention are described in the following paragraphs with reference to a receiver operating in accordance with the DVB-C2 standard, although it will be appreciated that embodiments of the present invention find application with other DVB standards and indeed other communications systems which utilise Orthogonal Frequency Division Multiplex (OFDM). 
     OFDM Data Transmission 
     OFDM systems transmit data simultaneously on a series of orthogonal frequency sub-carriers. The frequency sub-carriers transmitted over a period of time make up a series of OFDM symbols. 
     In order to reduce the effects of multi-path propagation and other effects, each OFDM symbol typically includes a guard interval. A guard interval is a portion of each OFDM symbol in the time domain that is copied from one end of the OFDM symbol and inserted at the other. Therefore the total duration of an OFDM symbol will be T=T g +T u , where T g  is the guard interval duration and T u  is the duration of useful data transmitted. The guard interval can determine the maximum length multi-path delay or simulcast delay which can be tolerated by the system. 
     A channel via which OFDM symbols are transmitted may vary in time and frequency. Therefore, each OFDM symbol may include pilot sub-carriers which are sub-carriers bearing pilot data of a known phase and amplitude. The pilot data from the pilot sub-carriers can be extracted at the receiver and enable a channel estimate of the channel to be derived. This allows the receiver to compensate for distortion added to the OFDM symbols by channel effects. 
     For the example of cable transmission, multi-path may be caused by several terminations being present on a cable leading to a partial reflection of the signal at each termination. 
     The distribution of pilot sub-carriers through a series of consecutive symbols may be arranged in accordance with a repeating pilot pattern. In some OFDM schemes, pilot patterns are defined by a variable, D x  which specifies the spacing between pilot sub-carriers across each symbol, and D y  which specifies how many OFDM symbols are required for each complete cycle of the repeating pattern. 
     In some OFDM systems, the consecutive series of OFDM symbols that are transmitted are arranged in a number of OFDM frames, each OFDM frame containing a number of OFDM symbols. In such systems, certain OFDM symbols may be designated preamble symbols which indicate the beginning of a frame and contain various signalling data. Such OFDM symbols may have a different pilot pattern than the payload bearing OFDM symbols of the frame. 
     OFDM Network 
       FIG. 1  provides an example block diagram of a typical OFDM network for transmitting data such as video images and audio signals in accordance with the DVB suite of OFDM standards such as DVB-T, DVB-H, DVB-T2 and in particular DVB-C2. In  FIG. 1  data is received by an encoder  101 . The encoder  101  encodes the data and then outputs the encoded data as a series of OFDM symbols. The encoding process includes applying interleaving and forward error correction to the data and then forming an OFDM symbol by inserting the data on the sub-carriers. The data includes pilot data as described above. Each OFDM symbol is then transformed into a time domain OFDM signal and a guard interval is added. The time domain OFDM signal is then output to a transmitter  102 . The transmitter  102  combines the time domain OFDM signal with a carrier signal forming a transmission signal and transmits this across a channel  103 . The nature of the channel  103  will depend on the type of OFDM standard being used and the infrastructure of the OFDM network. For example, the channel  103  may be a radio channel or may be a fix-wired data distribution network of a multimedia content provider. Once the transmission signal has been transmitted across the channel  103 , it is received by one or a plurality of receivers  104 . The receivers  104  will receive the transmitted signal in a manner depending on the nature of the network. For example, the receivers  104  may receive the transmission signal via a cable interface connected to the network of a multimedia content provider by a domestic telephone line or a link to a fibre optic network, or they may receive the transmitted signal using a radio signal reception interface. 
     OFDM Transmitter 
       FIG. 2  provides a more detailed diagram of the encoder  101 . The encoder  101  receives input data which includes signalling data and content data. The signalling data includes signalling information used to control the transmission of data across the network and provide information for the receivers  104  which is used when receiving and demodulating the transmitted OFDM symbols. The content data includes data relating to the content to be extracted at the receiver. This typically includes audio and video data but may include other data as well. In some examples, the content data can be divided into logical channels referred to as Physical Layer Pipes (PLPs). Each PLP can be encoded separately. 
     After being input to the encoder  101 , the content data is synchronised by an input synchroniser unit  202 . The input synchroniser unit  202  ensures a constant bit rate into the encoder  101  and compensates for end-to-end transmission delay for differing input data formats. The synchronised input data is then subject to error correction and interleaving by a forward error corrector and bit interleaving unit  203 . The data is then mapped onto an appropriate quadrature amplitude multiplex (QAM) constellation by a QAM mapper  204 . The various encoded PLP data stream are then formed into a series of data slices by a number of data slice builders  205 . The various data slices are then time and frequency interleaved by a plurality of time and frequency interleavers  206 ,  207  and constructed into a transmission frame by a frame builder  208  (the format of the transmission frame including data slices is explained further below). The transmission frame output from the frame builder  208  then has pilot data inserted, undergoes an inverse frequency transform to convert each OFDM symbol of the frame into a time domain signal and has a guard interval inserted by inverse frequency transform encoding block  209 . The output time domain signal then undergoes digital to analogue conversion by a digital to analogue converter  2010  and is output to the transmitter  102 . 
     The signalling data undergoes different encoding to the content data. Input signalling data is formed into layer 1 (L1) signalling data at the L1 signalling generator  2011 . The generated L1 signalling data is then subject to forward error correction and bit interleaving by an L1 forward error correction and bit interleaving unit  2012 . The bit interleaved and forward error corrected L1 signal is then time interleaved by time interleaver  2013  and formed into an L1 block by L1 block builder  2014 . The formation of the L1 block is explained in more detail below. The signalling data in the L1 block is then frequency interleaved by a frequency interleaver  2015  and enters the frame builder  208  to be combined into a frame along with the content data. 
     DVB-C2 Physical Layer Frame 
       FIG. 3  provides a schematic diagram illustrating the structure of a DVB-C2 physical layer frame which may be output from the frame builder  208 . As shown in  FIG. 3 , a DVB-C2 frame starts with L p  preamble OFDM symbols which are then followed by L Data  payload OFDM symbols. L Data  typically=448. Each OFDM symbol typically has  3408  active sub-carriers which can be allocated to pilot data, “data slices” and “notches”. 
     “Data slices” are groups of sub-carriers within each OFDM symbol derived from content data streams input into each data slice builder  205 . The number of active sub-carriers that can be allocated to a data slice is variable with a granularity determined by the pilot pattern being used i.e. a data slice width must be a multiple of the pilot sub-carrier spacing (D x ). 
     “Notches” are a number of adjacent sub-carriers within each OFDM symbol which are intended to accommodate for regions of the frequency spectrum that are likely to be subject to narrow band interference. Such interference may arise due to domestic power supplies and so on. Sub-carriers within a notch therefore do not carry any data that need be decoded by the receiver. 
     In DVB-C2, each preamble symbol typically has a sub-carrier pilot pattern of D x =6 and D y =1 whilst data symbols can have a sub-carrier pilot pattern with D x =12 and D y =4 or D x =24 or D y =4 corresponding to a guard interval of 1/64 or 1/128 respectively. Pilots are modulated by a PN sequence similar to DVB-T/T2. Each symbol has a duration Ts=Tu+Tg where Tu is the useful symbol duration and Tg is the guard interval duration being Tu/64 or Tu/128. 
       FIG. 4  provides a schematic diagram illustrating in more detail the process by which DVB-C2 preamble OFDM symbols are constructed within the encoder  101 . Each preamble symbol is composed of a  32  sub-carrier preamble header which carries some Layer 1 signalling information. All preamble symbols at the start of a given frame have identical headers.  FIG. 5  provides a schematic diagram of the L1 forward error correction and modulation unit  2012  which illustrates in more detail the way in which preamble headers may be generated. 
     Firstly, sixteen signalling bits from the L1 signalling generator  2011  are encoded by a Reed Muller encoder  501  with a half rate Reed-Muller (32, 16) code to give 32 bits. These bits are then processed in two ways. A first copy goes directly to a QPSK mapper  503  whilst the second copy goes through a 2-bit cycle left shifter  502 . This produces a sequence:
 
 u   i =λ(( i+ 2)mod 32) for  i= 0,1, . . . 31.
 
where λ, is the output sequence of the Reed-Muller encoder  501 . The sequence u is then scrambled with a 32 bit scrambling sequence w to provide the sequence ν. That is:
 
 νi=u   i   ⊕w   i  for  i= 0,1, . . . 31.
 
     The QPSK mapper  503  pairs up the bits of the sequences λ and ν such that λ provides the most significant bit and ν provides the least significant bit into QPSK symbol labels ready for modulation onto the first thirty two non-pilot sub-carriers of each preamble OFDM symbol of the current frame. 
     Receiver 
     Once the OFDM signal has been transmitted as a transmission signal via the channel  103  it is received by a receiver  104 .  FIG. 6  provides a schematic diagram illustrating the functional blocks of the receiver  104 . For the sake of brevity, only functional blocks pertinent to the present invention are discussed. However, it will be appreciated that the receiver will include other functional elements associated with conventional OFDM receivers such as channel estimation and correction units. 
     The receiver shown in  FIG. 6  includes a signal receive unit  601  for receiving the transmission signal. As explained above, the signal receive unit  601  will depend on the nature of the transmission signal. In some examples it may be a radio interface. In the example of DVB-C2 it may be a cable interface connecting to a cable service provider&#39;s data distribution network. 
     The received transmission signal is demodulated and sampled by demodulator and sampler  602  to extract a sampled time domain OFDM signal from the carrier signal. The time domain OFDM signal is then input to a symbol synchronisation detector  603  which is arranged to identify an optimum point for extracting samples of each OFDM symbol which have most useful energy in the time domain for triggering the Fourier transform. A symbol synchronisation time is thus generated. As will be explained below, the symbol synchronisation time is used to select a part of the OFDM time domain signal on which to apply a frequency transform operation. This is typically achieved by detecting the guard interval which separates each symbol. Once the symbol synchronisation detector  603  has identified the OFDM symbol boundaries in the time domain and generated a symbol synchronisation time, the symbol synchronisation time is further refined by a fine frequency offset detector  604 . The fine frequency offset detector  604  is arranged to remove any frequency offset errors up to +/−½ of the frequency width of a sub-carrier. Once the symbol synchronisation time has been processed and the fine frequency offset detector  604  used to correct for fine frequency offsets, the time domain OFDM signal is processed by a fast Fourier transform (FFT) processor  605 . The FFT processor  605  applies a frequency transform to a section of the time domain OFDM signal which converts this section back into the OFDM symbol in the frequency domain. The start point of the section, referred to as an FFT window, is determined using the symbol synchronisation time generated by the symbol synchronisation detector  602 . The transformed OFDM symbol is then processed by a coarse frequency offset detector  606 . The coarse frequency offset detector  606  is arranged to detect sub-carrier positional frequency offset errors which are a multiple of the frequency width of a sub-carrier. The coarse frequency offset detector  606  can be arranged to detect such frequency offset errors by using the position of pilot sub-carriers in the OFDM symbol. 
     When the OFDM symbol has been corrected for coarse frequency errors, the OFDM symbol is de-interleaved by a frequency de-interleaver  607  and then processed by a frame synchronisation detector  608  arranged to detect boundaries between OFDM frames and thus the position of the OFDM symbol within the OFDM frame. Once the frame synchronisation has been established, data can be extracted from the sub-carriers of the OFDM symbol in accordance with the frame structure. This is undertaken by various components but is represented in  FIG. 6  generally by a data extractor  609 . Once the data has been extracted it can be used appropriately in the receiver. For example, signalling data can be used to provide further information necessary for the processing of the received data and content data can be decoded and output, for example as audio and video signals 
     Various parts within the receiver  104  will now be explained in more detail with reference to  FIGS. 7 to 17 . 
     Symbol Synchronisation Detector 
     As mentioned above, a symbol synchronisation time must be determined to position the start of the FFT window. In order to do this, the OFDM symbol boundaries (i.e. a position in the time domain OFDM signal from which an optimum frequency transform can be applied) must be identified in the time domain OFDM signal. In the time domain, each OFDM symbol is preceded by a guard interval which is filled with a copy of the last Ng samples of the OFDM symbol—a so-called cyclic prefix. Symbol boundaries can therefore be acquired from the time domain OFDM signal by identifying the presence of a guard interval. Since the samples of the guard interval appear twice for each OFDM symbol, a correlator can be used to identify the position of the guard interval in the time domain OFDM signal. In DVB-C2, OFDM symbols have one of two possible guard lengths. The symbol synchronisation detector  603  therefore has two tasks: to find the duration of the guard interval; and to determine an optimal symbol synchronisation time to begin the FFT window. 
       FIG. 7  provides a schematic diagram showing a symbol synchronisation detector  603  arranged in accordance with an embodiment of the present invention. The symbol synchronisation detector  603  comprises two correlators. The first correlator  6031  is arranged to correlate the time domain OFDM signal in accordance with a guard interval of 1/64. In other words, the first correlator  6031  is arranged to look for an auto correlation within the OFDM time domain signal of length T u /64 indicating the presence of a 1/64 guard interval. The second correlator  6032  is arranged to correlate the time domain OFDM signal in accordance with a guard interval of 1/128. In other words, the second correlator  6032  is arranged to look for an auto correlation within the OFDM time domain signal of length T u /128 indicating the presence of a 1/128 guard interval. The first and second correlators run concurrently in parallel. 
     The symbol synchronisation detector  603  shown in  FIG. 7  is arranged in accordance with the DVB-C2 standard which as mentioned above provides only two possible guard lengths. However, it will be appreciated that the symbol synchronisation detector  603  could be arranged with more correlators for an OFDM scheme employing more than two guard interval lengths. 
     Each correlator includes a matching filter  6034 ,  6036  and accumulator memory  6035 ,  6037 . The matching filter  6034  of the first correlator is arranged to output a signal which increases in level the more closely sections of the input OFDM signal of a length T u /64, matches with itself. The accumulator memory  6035  accumulates the output of the matching filter. The operation of the matching filter  6035  and the accumulator memory  6037  from the second correlator  6032  run in parallel to this and are the same except that the matching filter  6036  outputs a signal which increases in size the more closely sections of the input OFDM signal of a length T u /128, matches with itself. 
     The initial input sequence of samples r(n) from the demodulator and sampler  602  is input to a complex conjugator  6040  and a first delay  6044 . The output of the complex conjugator  6040  and the delay  6044  are multiplied by a multiplier  6044  forming an output stream a(n). The sequence a(n) is input to each matching filter  6032 ,  6034  where it is delayed by a second delay  6046 , and subtracted with a(n) by an adder  6048  and a delayed version of a(n) provided by a third delay  6050 . The output of the matched filter  6032  is then designated either A(n) or B(n) for the first and second matched filters respectively. 
     Specifically, the first correlator  6031  implements the following equation (where * denotes a complex conjugation operation): 
     
       
         
           
             
               
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               where 
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     And the second correlator  6032  implements the following equation: 
     
       
         
           
             
               
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                 m 
               
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                 n 
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               where 
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     The accumulator memories  6035 ,  6037  then execute an element-by-element accumulation of the vectors A m (n) and B m (n) for n=0, 1, 2 . . . N s , where N s =4128 for the first correlator  6031 ; and N S =4160 for the second correlator  6032 ) and m is the OFDM symbol index. 
       FIG. 8  provides a schematic diagram indicating idealised traces of the signals r(n), a(n) and A(n) that flow at particular points in symbol synchronisation detection unit  603  shown in  FIG. 7 . 
     Returning to  FIG. 7 , the accumulator memories  6035  and  6037  of the first and second correlators are monitored by a correlation detection processor  6033 . Because the output of each matching filter increases in level when it detects the guard interval length it is arranged to detect, the accumulator memory storing the greatest value indicates the length of the guard interval for a given received OFDM time domain signal. For example if the accumulator memory  6035  of the first correlator  6031  has the highest value after a set number of symbol times, this indicates that the OFDM time domain signal has a guard interval of T u /128. 
     In some examples, the correlation detect processor  6033  may wait for a given period of time and then determine the guard interval based on the accumulator memory with the highest value. Alternatively, the correlation detect processor  6033  may instead wait until one of the accumulator memories reaches a threshold value. 
     In one example, after a given number L sync  OFDM symbols (each of length Ns=4160 in the example of DVB-C2), the correlation detect processor  6033  detects which accumulator memory gives a higher peak correlation value. The guard interval length Ng is then decided: 
     
       
         
           
             
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     In another example, both correlators are run until either max(A) or max(B) exceeds a given threshold. Then the corresponding N g  (i.e. guard interval length) is chosen. The corresponding accumulator memory is then reset and the selected branch run again for L sync  OFDM symbols. 
     The correlation detect processor  6033  is arranged to use the time at which the guard interval of the determined length has been detected to determine the symbol synchronisation time that the FFT processor  605  can use to set the start time of the FFT window. 
     In one example, the symbol start time τ can be derived by implementing the following equation if the second accumulator memory  6037  has the greater value:
 
τ=Index(max( B ))mod  N   s .
 
     Or, if the first accumulator memory  6035  is detected as having the greater value, the correlation detector processor  6033  resets the first accumulator memory  6035  and controls the first correlator  6031  to run for another L sync  symbols of length N s =4128. The symbol synch time τ is then calculated as being
 
τ=Index(max( A ))mod  N   s .
 
     Accordingly the correlation detect processor  6033  is arranged to output the symbol synch time τ. 
       FIG. 9   a  provides an illustration of two graphs indicating the result of the first stage of correlation for the first correlator  6031  and the second correlator  6032  in which max(A)&gt;max(B) after averaging over 10 OFDM symbols. This is for a transmission system using the 1/128 guard interval. 
       FIG. 9   b  provides an illustration of two graphs indicating values of the contents of the first accumulator memory  6035  after reset and subsequent averaging over 10 OFDM symbols. 
     Fine Frequency Offset 
     As mentioned above, the fine frequency offset detector  604  is arranged to determine a fine frequency offset error Ω. In some examples this can be determined by implementing the following equation:
 
Ω=arg(max( A ))
 
     if the first accumulator memory  6035  is detected as having the greater value, or
 
Ω=arg(max( B ))
 
     if the second accumulator memory  6037  is detected as having the greater value. 
     Accordingly, the output of the correlation detect processor  6033  in the symbol synchronisation detector which is input into the fine frequency offset detector  604  includes max(A) or max(B) depending on which guard length interval is detected. 
     Coarse Frequency Offset 
     As described above, the guard interval correlation used to acquire the OFDM symbol boundary also provides the fine (i.e. sub-FFT bin) frequency offset. This precludes frequency offsets that are a multiple of one FFT bin (i.e. sub-carrier width in frequency domain). In other words, the sub-carriers of the OFDM symbol may be displaced in the frequency domain by one or more sub-carrier positions with respect to an original position within the OFDM symbol which they occupied prior to transmission. Such displacement amounts to a “coarse” frequency offset of the sub-carriers of the frequency transformed OFDM symbol and may arise for various reasons such as errors in the de-modulation process or errors imposed on the transmission signal by channel conditions. In the presence of such frequency offsets, data on a sub-carrier of the OFDM symbol originally transmitted in bin number (i.e. sub-carrier number) k would actually be located in bin number k±ω where ω is the multiple and the actual frequency offset being ωΔf where Δf in Hertz is the bandwidth of each FFT bin. Thus ω has to be estimated in order determine the coarse frequency offset. 
     In one example in accordance with DVB-C2, for each preamble symbol, between the upper or lower edge of a tuned rasta channel and the first useful OFDM sub-carrier,  344  (i.e. (4096−3408)/2) sub-carriers are set to zero at the transmitter. This means that potentially a coarse frequency offset of ω= 344  can be accommodated. In each DVB-C2 preamble symbol, pilot sub-carriers occur every 6th sub-carrier. The pilot sub-carriers are modulated with a unique random sequence known both at the transmitter and the receiver. After performing the FFT on the preamble symbol in the receiver, this sequence can be located within the OFDM symbol by matching the expected pilot sub-carrier sequence with sub-carriers at the appropriate locations in the frequency spectrum.  FIG. 10  illustrates a technique to detect the coarse frequency offset. 
     In  FIG. 10 , a first row of circles  1010  represents an OFDM symbol comprising sub-carriers with a maximum positive offset. The shaded circles  1011  represent preamble pilot sub-carriers whilst the un-shaded circles  1012  represent non-pilot sub-carriers. A second row of circles  1020  represents an OFDM symbol of sub-carriers with a zero offset. A third row of circles  1012  represents an OFDM symbol of sub-carriers with a maximum negative offset. The maximum positive and negative offsets are parameters that represent a maximum value of ω that is expected. 
     In order to determine the coarse frequency offset once the OFDM signal has undergone the FFT and the OFDM symbol has been recovered in the frequency domain, all of the OFDM symbol sub-carriers are shifted from right to left through a pilot data filter. The output of the pilot data filter will then peak when the position of the pilot sub-carriers is detected. Accordingly the coarse frequency offset can be calculated. 
       FIG. 11  provides a schematic diagram showing an implementation of the coarse frequency offset detector  606  arranged to perform this technique. The input of the coarse frequency offset detector is the OFDM symbol recovered from the OFDM time domain signal by the FFT processor  605 . The coarse frequency offset detector  606  includes a pilot data filter  6061 . The pilot data filter  6061  has a delay stage  6062  for each pilot sub-carrier. As mentioned above, the pilot sub-carriers of the preamble symbol are modulated with a unique random sequence known both at the transmitter and the receiver. The output taps of each delay stage are scaled with a value (P 0  to P 570  for DVB-C2) corresponding to this random sequence. The output of these combinations are then summed by a summation unit  6063  within the pilot data filter  6061 . A pulse detector  6064  monitors the output of the pilot data filter  6061 . When a pulse is detected, the unique sequence of pilot sub-carriers is detected. A signal is then sent to a coarse frequency detect processor which determines from the time of the pulse a value for the coarse frequency offset ω. 
     In some examples, the output of the pulse detector is only valid between offsets of 3408+(343−Max Offset) and 4096−(344−Max Offset). Thus if the maximum expected frequency offset is ±200Δf i.e. Max Offset=200, then the pulse detector output will be valid from shift  3551  to shift  3952  i.e. (3952−3551)=a total offset of 401 FFT bins. If a shift of an FFT bin is numbered from −Max Offset to +Max Offset then the pulse detector output will go high for the shift that corresponds to the observed frequency offset. 
       FIG. 12  provides an example plot of the input of the pulse detector  6064  for a frequency offset of ω=−30 FFT bins in a case where the Max Offset is set to 100. In some examples the pulse detector  6064  might use a threshold to clip this input as a detection of the presence (or absence) of a substantial pulse. 
     Frame Acquisition 
     As explained above, after correcting for the coarse frequency offset, the next stage is frame acquisition undertaken by the frame synchronisation detector  608 . The function of the frame synchronisation detector  608  is to find the boundary of each OFDM frame by detecting which of the received OFDM symbols are preamble OFDM symbols found at the beginning of each OFDM frame. 
     The detection of the coarse frequency offset detailed above is based on the detection of pilot sub-carriers from the preamble OFDM symbol. It is therefore known that the OFDM symbol on which the frequency offset is detected is a preamble OFDM symbol. However, if there are more than one preamble OFDM symbols per frame (i.e. Lp&gt;1) then the relative position of this preamble OFDM symbol in the set of preamble OFDM symbols of the current frame is not known. Accordingly, frame acquisition cannot be derived directly from the process of acquiring the coarse frequency offset value ω. 
     Preamble Pilot Correlation 
     In a first example, the beginning of a frame can be detected by attempting to identify the first occurrence of a preamble pilot sub-carrier correlation corresponding to the frame boundary. Therefore, in a similar way to the frequency offset estimation implemented in the coarse frequency offset detector  606 , preamble pilot sub-carrier correlation can be used to acquire the start of the OFDM frame. In this case however, only the zero-th tap of the pilot data filter need be used. The presence of a pulse at the output will indicate that the current symbol is a preamble symbol and the frame is identified as beginning when the first pulse is detected (corresponding to the first preamble symbol). The preamble header can then be decoded to extract the L1_INFO_SIZE signalling data and L1_T1_MODE signalling data prior to decoding of the Layer 1 part 2 information. In this example, the function of the frame synchronisation detector  608  can be performed within the coarse frequency offset detector  606 . 
       FIG. 13  provides an illustration of a graph of the input of the pulse detector  6064  for a system in which there are twenty symbols (Lf=20) in a frame and there are four (Lp=4) preamble symbols per frame both in a clear channel and with a signal to noise ratio of 6 dB. 
     Preamble Header Correlation 
     In the above example, the beginning of the frame is detected by correlating the known values of the pilot sub-carriers from the preamble OFDM symbol with the received OFDM symbols. However, in another example it is possible to identify the beginning of the frame by identifying the preamble header sequence which is inserted into each preamble OFDM symbol of the frame. As mentioned above, this header sequence is unique to the preamble symbols in each frame. 
     As explained above,  FIGS. 4 and 5  illustrate how the preamble header is constructed and encoded. At the encoder, data to be inserted on the first thirty two sub-carriers of the preamble symbol which carries the header is shuffled with the other L1 part 2 data in a frequency interleaver prior to pilot data insertion. However, in the receiver after each OFDM symbol has been frequency de-interleaved by the frequency de-interleaver  607  the header data is returned to its original position on the first thirty two sub-carriers of the preamble OFDM symbol. Therefore, in one example, preamble header correlation can be achieved by extracting the relevant data from the output of the frequency de-interleaver (i.e. the first thirty two sub-carriers from every OFDM symbol) and reversing the process undertaken by the circuit shown in  FIG. 5  to locate the occurrence of the preamble header. Once this has been identified, the frame boundary has been found. 
     Hard Decision Matching 
       FIG. 14   a  provides a schematic diagram illustrating a first implementation of the frame synchronisation detector  608  using the header correlation technique described above. The frame synchronisation detector  608  shown in  FIG. 14   a  implements a “hard-decoding” algorithm to detect the preamble header of the OFDM frame (i.e. where absolute binary values are processed to detect the preamble header). 
     The frame synchronisation detector  608  receives de-interleaved OFDM symbols from the symbol de-interleaver  607 . These are then input to a data extractor  6081  which extracts the QPSK data carried on the first thirty two sub-carriers of the symbol, R(k) (k=0, 1, . . . 31). This provides an output R(k) which is input to a QPSK de-multiplexer  6082 . As shown in  FIG. 5 , during the construction of the preamble header, two sequences λ and ν are mapped into QPSK labels such that λ provides a sequence corresponding to the most significant bits of the QPSK labels and ν provides the least significant bits of the labels. For example, if λ={0, 1, 1, 0, . . . } and ν={0, 0, 1, 1, . . . } the resultant QPSK cell labels would be {00, 10, 11, 01, . . . }. 
     The QPSK de-multiplexer  6082  divides the QPSK data into a first stream λ′ providing a sequence corresponding to the most significant bits of the demodulated QPSK data, and second stream ν′ providing the least significant bits of the de-modulated QPSK data. The bits comprising the second stream ν′ undergo an XOR operation with a 32 bit MPS scrambling sequence w provided by a MPS sequence generator  6084  and corresponding to the scrambling sequence used at the encoder. The result of this is a modified stream u′ which is then cyclically right shifted by two bits by a cyclic shifter  6085 . This provides a further modified stream u″. 
     The first stream λ′ and the modified stream u″ are then compared in a Hamming distance calculator  6086  to determine a Hamming distance between the two streams. The output of the Hamming distance calculator  6086 , Δ, is then subtracted from thirty two by a subtractor  6087 . The output of the subtractor is fed into a pulse detector  6089 . When the Hamming distance Δ of u″ and λ′ is at a minimum (indicating that the data from the first thirty two sub-carriers corresponds to the preamble header and the beginning of an OFDM frame) the pulse detector  6089  detects a pulse at the output of the subtractor  6087 . The output of the pulse detector is input to a frame synchronisation detect processor  60810  which processes the output of the pulse detector and outputs a signal f t  indicating the that beginning of a frame has been detected. 
     An equivalent procedure is to leave ν′ unchanged, cyclic left shift λ′, scramble the result with the 32 bit MPS sequence w to give λ″. The Hamming distance Δ can then be computed between v′ and λ″.  FIG. 14   b  provides a schematic diagram illustrating this alternative implementation of the frame synchronisation detector  608 . 
       FIG. 15  provides a diagram illustrating a graph providing an indication of the output of the pulse detector  6089  using the frame synchronisation technique implemented in the frame synchronisation detectors shown in  FIGS. 14   a  and  14   b . In particular, the graph shown in  FIG. 15  shows a series of peaks  7001 ,  7002  indicating a frame has been detected. A frequency of the peaks indicates a corresponding frequency in the occurrence of frames. For example, the graph of  FIG. 15  indicates a frame rate of one frame per twenty symbols. 
     Soft Decision Matching 
       FIG. 16   a  provides a schematic diagram illustrating another implementation of the frame synchronisation detector  608 . The frame synchronisation detector shown in  FIG. 16   a  a implements a “soft-decoding” algorithm to detect the preamble header of the OFDM frame (i.e. where continuous rather than absolute binary values are processed to detect the preamble header). 
     In the implementation of the frame synchronisation detector shown in  FIG. 16   a , the data stream R(k) representing data extracted from the first thirty two sub-carriers of the de-interleaved OFDM symbol are split into a real component stream λ′(k)=real(R(k)) and an imaginary component stream ν′(k)=imag(R(k)) by a QPSK slicer  60812 . The MPS scramble sequence w is converted into a bipolar form by a bipolar converter  60814  to produce scrambling signal w′(k) in bipolar form (i.e. w′(k)=1−2w(k)). W′(k) is then multiplied with v′(k) to produce a signal u′(k) which is cyclically shifted by two to produce a signal u″(k). u″(k) and λ′(k) are then multiplied and summed by a summation unit  60813 . The output of the summation unit is connected to a pulse detector  6089 . The input to the pulse detector can be represented by the following equation: 
     
       
         
           
             A 
             = 
             
               
                 ∑ 
                 
                   k 
                   = 
                   0 
                 
                 31 
               
               ⁢ 
               
                   
               
               ⁢ 
               
                 
                   
                     u 
                     ″ 
                   
                   ⁡ 
                   
                     ( 
                     k 
                     ) 
                   
                 
                 * 
                 
                   
                     λ 
                     ′ 
                   
                   ⁡ 
                   
                     ( 
                     k 
                     ) 
                   
                 
               
             
           
         
       
     
     The pulse detector  6089  operates in conjunction with a frame synchronisation detector  60810  in generally the same way as described above with reference to  FIG. 14   a . Accordingly when the summation unit  60813  outputs a peak for a given symbol, this indicates that the current symbol being processed contains a preamble header and thus represents the beginning of an OFDM frame. 
     In a similar fashion to the hard decision implementation, an equivalent procedure is to leave ν′ unchanged, cyclic left shift λ′, scramble the result with the 32 bit MPS sequence w to give k″. Then input to the pulse detector can be represented by the following equation: 
     
       
         
           
             A 
             = 
             
               
                 ∑ 
                 
                   k 
                   = 
                   0 
                 
                 31 
               
               ⁢ 
               
                   
               
               ⁢ 
               
                 
                   
                     v 
                     ′ 
                   
                   ⁡ 
                   
                     ( 
                     k 
                     ) 
                   
                 
                 * 
                 
                   
                     λ 
                     ″ 
                   
                   ⁡ 
                   
                     ( 
                     k 
                     ) 
                   
                 
               
             
           
         
       
     
     This implementation is shown in  FIG. 16   b.    
       FIG. 17  provides a schematic diagram showing a graph providing an indication of the output of the pulse detector  6089  using the frame synchronisation technique implemented in the frame synchronisation detectors shown in  FIGS. 16   a  and  16   b . In particular, the graph shown in  FIG. 17  shows a series of peaks  8001 ,  8002  indicating a frame has been detected. A frequency of the peaks indicates a corresponding frequency in the occurrence of frames. For example the graph of  FIG. 17  indicates a frame rate of one frame per twenty symbols. 
     Summary of Operation 
       FIG. 18  provides a flow diagram illustrating a process undertaken in accordance with an embodiment of the present invention. 
     At a first step S 101 , a signal representing the OFDM symbols is detected. At a second step S 102  a sampled version of the OFDM symbols in the time domain is generated. At third step S 103 , each OFDM symbol is concurrently auto correlated using a plurality of correlation processes, each correlation process auto correlating the OFDM symbol with a length of samples corresponding to one of the plurality of predetermined lengths and the time domain start point of each symbol is determined. At a fourth step S 104  a frequency transform is applied to the sampled OFDM symbol based on the time domain start point. At step S 105  the coarse frequency offset is calculated. 
     Various modifications may be made to the embodiments herein before described. For example it will be understood that the particular component parts of which the receiver described above is comprised, for example the symbol synchronisation detector, the fine frequency offset detector, the coarse frequency offset detector and the frame synchronisation detector are essentially logical designations. Accordingly, the functionality that these component parts provide may be manifested in ways that do no conform precisely to the forms described above and shown in the diagrams. For example aspects of the invention be implemented in the form of a computer program product comprising instructions that may be implemented on a processor stored on a data carrier such as a floppy disk, optical disk, hard disk, PROM, RAM, flash memory or any combination of these or other storage media, or transmitted via data signals on a network such as an Ethernet, a wireless network, the Internet, or any combination of these of other networks, or realised in hardware as an ASIC (application specific integrated circuit) or an FPGA (field programmable gate array) or other configurable or bespoke circuit suitable to use in adapting the conventional equivalent device. 
     It will be appreciated that the present invention is not limited to application with DVB and may be extended to other standards for transmission or reception, both fixed and mobile.