Patent Publication Number: US-8994343-B2

Title: Switching power supply circuit, and method for control of switching power supply circuit

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is the National Phase of International Application No. PCT/JP2011/057185, filed on Mar. 24, 2011, which claims priority under 35 U.S.C. 119(a) to Patent Application No. 2010-071916, filed in Japan on Mar. 26, 2010. 
     TECHNICAL FIELD 
     The present invention relates to a switching power supply circuit and a method for control of a switching power supply circuit, and particularly relates to a power factor correction circuit. 
     BACKGROUND ART 
     As a power factor correction circuit for correcting an input-side power factor, a circuit (so-called booster circuit) including a reactor, a diode, and a switching element has been conventionally proposed. More specifically, the reactor and the switching element are connected in series with each other between two input terminals, and the diode and the switching element are connected in series with each other between two output terminals. The diode is provided with the anode thereof directed to the switching element side. A smoothing capacitor is provided between the two output terminals. 
     In this circuit, when the switching element is conducting, a current flows in the input terminals via the reactor and the switching element, and when the switching element is non-conducting, a current flows in the input terminals via the reactor, the diode, and the output terminals. This increases the conducting angle of an input current, and thereby corrects an input-side power factor. 
     A power factor correction circuit including two such circuits has been also proposed. In such a power factor correction circuit, switching elements belonging to the two circuits are rendered conducting at different timings. This power factor correction circuit is referred to as a so-called interleave type power factor correction circuit. 
     Moreover, Japanese Patent Application Laid-Open No. 11-289766 (1999) is disclosed as a technique related to the present invention. 
     SUMMARY OF THE INVENTION 
     Problems to be Solved by the Invention 
     In the switching power supply circuit including two circuits mentioned above, there is still room for innovation from the viewpoint of electrical characteristics of the switching power supply circuit in accordance with a variation of a load of the smoothing capacitor in a case where the load varies. 
     Therefore, an object of the present invention is to provide a switching power supply circuit that contributes to achievement of electrical characteristics in accordance with a variation of a load. 
     Means for Solving the Problems 
     A first aspect of a switching power supply circuit according to the present invention includes: first and second input terminals (P 1 , P 2 ); first and second output terminals (P 3 , P 4 ); a second power supply line (LL) connecting the second input terminal and the second output terminal to each other; and a plurality of circuits ( 3 ,  3   a ,  3   b ), wherein each of the plurality of circuits includes: a first power supply line (LH 1 , LH 2 ) connecting the first input terminal and the first output terminal to each other; a reactor (L 1 , L 2 ) provided on the first power supply line; a diode (D 11 , D 21 ) connected in series with the reactor on the first power supply line and arranged with an anode thereof directed toward the reactor; and a switching element (S 1 , S 2 ) provided between the second power supply line and a point between the reactor and the diode, and wherein characteristics of the reactor of one of the plurality of circuits and the reactor of another of the plurality of circuits are different from each other, or characteristics of the switching element of the one of the plurality of circuits and the switching element of the another of the plurality of circuits are different from each other, or characteristics of the diode of the one of the plurality of circuits and the diode of the another of the plurality of circuits are different from each other. 
     A second aspect of a switching power supply circuit according to the present invention is the switching power supply circuit according to the first aspect, wherein the switching element (S 1 ) belonging to the one of the plurality of circuits ( 3   a ) is an insulated gate bipolar transistor, and the switching element (S 2 ) belonging to the another of the plurality of circuits ( 3   b ) is a MOS field effect transistor. 
     A third aspect of a switching power supply circuit according to the present invention is the switching power supply circuit according to the first aspect, wherein the switching element (S 1 ) belonging to the one of the plurality of circuits ( 3   a ) is formed of a silicon carbide semiconductor or a gallium nitride semiconductor, and the switching element (S 2 ) belonging to the another of the plurality of circuits ( 3   b ) is formed of a silicon semiconductor. 
     A fourth aspect of a switching power supply circuit according to the present invention is the switching power supply circuit according to any one of the first to third aspects, wherein an impedance of the reactor (L 1 ) belonging to the one of the plurality of circuits ( 3   a ) is lower than an impedance of the reactor belonging to the another of the plurality of circuits ( 3   b ). 
     A first aspect of a method for control of a switching power supply circuit according to the present invention is a method for control of the switching power supply circuit according to any one of the first to fourth aspects, the method including performing: a first step of keeping the switching element (S 1 , S 2 ) belonging to each of the plurality of circuits ( 3   a ,  3   b ) non-conducting; and a second step of when a current flowing in the first and second input terminals (P 1 , P 2 ) exceeds a first predetermined value (Iref 1 ), repeatedly switching conducting/non-conducting of the switching element (S 1 ) belonging to the one of the plurality of circuits ( 3   a ). 
     A second aspect of a method for control of a switching power supply circuit according to the present invention is the method for control of the switching power supply circuit according to the first aspect, wherein in the second step, the conducting/non-conducting of the switching element (S 1 ) belonging to the one of the plurality of circuits ( 3   a ) is repeatedly switched based on a first DC voltage command value that is higher than a voltage between the first and second output terminals (P 3 , P 4 ) in the first step. 
     A third aspect of a method for control of a switching power supply circuit according to the present invention is the method for control of the switching power supply circuit according to the first or second aspect, further including performing a third step of, when the current flowing in the first and second input terminals (P 1 , P 2 ) exceeds a second predetermined value (Iref 2 ) that is higher than the first predetermined value (Iref 1 ), repeatedly switching conducting/non-conducting of the switching element (S 1 ) of the one of the plurality of circuits ( 3   a ) and the switching element (S 2 ) of the another of the plurality of circuits ( 3   b ). 
     A fourth aspect of a method for control of a switching power supply circuit according to the present invention is the method for control of the switching power supply circuit according to the third aspect, wherein in the third step, the conducting/non-conducting of the switching element (S 1 ) of the one of the plurality of circuits ( 3   a ) and the switching element (S 2 ) of the another of the plurality of circuits ( 3   b ) is repeatedly switched based on a second DC voltage command value that is higher than the first DC voltage command value. 
     A fifth aspect of a method for control of a switching power supply circuit according to the present invention is the method for control of the switching power supply circuit according to any one of the first to fourth aspects, further including, prior to the first step and the second step, performing: a fourth step of keeping the switching element (S 1 , S 2 ) belonging to each of the plurality of circuits ( 3 ,  3   a ,  3   b ) non-conducting, and obtaining a first relationship between the current and an efficiency of the switching power supply circuit; and a fifth step of repeatedly switching conducting/non-conducting of the switching element (S 1 ) of the one of the plurality of circuits ( 3   a ) based on the first DC voltage command value, and obtaining a second relationship between the current and an efficiency of the switching power supply circuit, wherein the current that gives the same efficiency both in the first relationship and in the second relationship is adopted as the first predetermined value (Iref 1 ). 
     Effects of the Invention 
     As for the first aspect of the switching power supply circuit according to the present invention, a description will be given below with an example case where the plurality of circuits are two first and second circuits. 
     The characteristics of at least one of the reactor, the switching element, and the diode of the first circuit are different from the characteristics of at least one of the reactor, the switching element, and the diode of the second circuit. Accordingly, the electrical characteristics of the switching power supply circuit in a first state where only the first circuit is operated are different from those in a second state where only the second circuit is operated. The electrical characteristics of the switching power supply circuit in a third state where both of the first and second circuits are operated are different from the electrical characteristics in the first state and in the second state. 
     As described above, three types of electrical characteristics, as the switching power supply circuit, can be exhibited with the two first and second circuits. 
     Moreover, the electrical characteristics in the first to third states influence the power factor, the efficiency, and harmonics, respectively. Therefore, by appropriately selecting the first to third states, the power factor, the efficiency, and harmonics can be appropriately adjusted. 
     In this switching power supply circuit, for example, the first to third states can be appropriately selected in accordance with a variation of a load connected to the first and second output terminals. Accordingly, this contributes to achievement of the electrical characteristics in accordance with the variation of the load. 
     In the second aspect of the switching power supply circuit according to the present invention, the circuit having a MOS field effect transistor has a smaller current capacity than that of an insulated gate bipolar transistor, and is widely used in a television receiver or the like. This allows the use of a general-purpose component, and thus a cost is reduced. 
     In the third aspect of the switching power supply circuit according to the present invention, the switching element formed of a silicon carbide semiconductor or a gallium nitride semiconductor has a lower conduction loss and a higher voltage resistance than those of the switching element formed of a silicon semiconductor. 
     In the fourth aspect of the switching power supply circuit according to the present invention, the reactor belonging to the one of the plurality of circuits can be downsized. 
     In the first aspect of method for control of the switching power supply circuit according to the present invention, when the current is lower than the first predetermined value, all the switching elements are rendered non-conducting. In a state where the current is low, a switching loss occupies a large proportion of a loss occurring in the circuit, and therefore the efficiency in a state where the current is lower than the first predetermined value can be enhanced. On the other hand, when the current is higher than the first predetermined value, the conducting/non-conducting of the switching element belonging to the one of the plurality of circuits is repeatedly switched, and thereby the efficiency in this state can be improved. 
     In the second aspect of the method for control of the switching power supply circuit according to the present invention, a stable operation of the circuit in the second step is achieved in a range where the current is high, as compared with a control method in which the first DC voltage command value is set equal to the DC voltage in the first step. On the other hand, the efficiency of the circuit in a range where the current is low can be enhanced. 
     In the third aspect of the method for control of the switching power supply circuit according to the present invention, in a state where the current is low, the switching loss occupies a large proportion of the loss occurring in the circuit, and therefore the efficiency in a state where the current is lower than the second predetermined value can be enhanced. On the other hand, when the current is higher than the second predetermined value, the conducting/non-conducting of the switching elements belonging to the one and the another of the plurality of circuits is repeatedly switched, and thereby the efficiency in this state can be improved. 
     In the fourth aspect of the method for control of the switching power supply circuit according to the present invention, the efficiency in the third step can be improved, as compared with a control method in which the second DC voltage command value is set equal to the first DC voltage command value. 
     In the fifth aspect of the method for control of the switching power supply circuit according to the present invention, the first step and the second step can be selected so as to obtain the highest efficiency. 
     These and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIGS. 1 and 2  are diagrams showing one example of a conceptual configuration of a switching power supply circuit; 
         FIG. 3  is a conceptual diagram showing one example of a state of conducting/non-conducting of each switching element, currents flowing in reactors, and the sum of the currents flowing in the reactors; 
         FIG. 4  is a diagram showing one example of a conceptual configuration of the switching power supply circuit; 
         FIG. 5  is a diagram showing a time period during which each outside air temperature occurs in Japan, and the relationship between the outside air temperature and an air conditioning load; 
         FIGS. 6 to 8  are diagrams schematically showing one example of the relationship of the efficiency relative to a current; 
         FIGS. 9 to 11  are conceptual diagrams showing one example of currents flowing in the reactors and the sum thereof; and 
         FIG. 12  is a conceptual diagram showing one example of currents. 
     
    
    
     EMBODIMENT FOR CARRYING OUT THE INVENTION 
     First Embodiment 
     As illustrated in  FIG. 1 , a switching power supply circuit includes a plurality of circuits  3 , input terminals P 1 , P 2 , and output terminals P 3 , P 4 . 
     A DC voltage is applied between the input terminals P 1 , P 2 . In the illustration of  FIG. 1 , a diode rectifier circuit  2  is connected to the input terminals P 1 , P 2 . The diode rectifier circuit  2  rectifies an AC voltage supplied from an AC power supply  1 , and applies a DC voltage obtained after the rectification between the input terminals P 1 , P 2 . Here, a potential applied to the input terminal P 2  is lower than a potential applied to the input terminal P 1 . It is not an essential requisite that the diode rectifier circuit  2  is connected to the input terminals P 1 , P 2 . It suffices that any configuration for applying a DC voltage between the input terminals P 1 , P 2  is connected to the input terminals P 1 , P 2 . 
     Each of the plurality of circuits  3  is connected to the input terminals P 1 , P 2  and to the output terminals P 3 , P 4 . Each of the circuits  3  functions as a booster circuit to boost the DC voltage applied to the input terminals P 1 , P 2  and also functions as a power factor correction circuit to improve an input-side power factor, as will be described later. 
     A smoothing capacitor C 1  is provided between the output terminals P 3 , P 4 . The smoothing capacitor C 1  smoothes the DC voltage boosted by each circuit  3 . 
     A capacitor C 2  may be provided between the input terminals P 1 , P 2 . The capacitor C 2  can reduce a noise of a current inputted to each circuit  3 . 
       FIG. 2  conceptually shows one example of a specific configuration of the plurality of circuits  3 . In  FIG. 2 , as an example, two circuits  3   a ,  3   b  are shown. In the illustration of  FIG. 2 , outputs of the circuits  3   a ,  3   b  are inputted to an inverter  4  via the smoothing capacitor C 1 . That is, the output terminals P 3 , P 4  are connected to the inverter  4  at the input side of the inverter  4 . 
     The input terminal P 2  and the output terminal P 4  are connected to each other by a power supply line LL. 
     The circuit  3   a  includes a power supply line LH 1 , a reactor L 1 , a diode D 11 , and a switching element S 1 . The power supply line LH 1  connects the input terminal P 1  and the output terminal P 3  to each other. The reactor L 1  is provided on the power supply line LH 1 . The diode D 11  is connected in series with the reactor L 1  at the output terminal P 3  side of the reactor L 1 . The diode D 11  is provided with the anode thereof directed to the reactor L 1 . 
     The switching element S 1  is provided between the power supply line LL and a point between the reactor L 1  and the diode D 11 . The conducting/non-conducting of the switching element S 1  is controlled by the control section  5 . In the illustration of  FIG. 2 , the switching element S 1  includes a transistor T 1  and a diode D 12 . The transistor T 1  is, for example, an insulated gate bipolar transistor, and provided with an emitter electrode thereof directed to the power supply line LL side. It is not always necessary that the switching element S 1  includes the transistor T 1  and the diode D 12 . For example, it may be acceptable that the diode D 12  is not provided. As the switching element S 1 , for example, a MOS (Metal-Oxide-Semiconductor) field effect transistor may be adopted. 
     The circuit  3   b  includes a power supply line LH 2 , a reactor L 2 , a diode D 21 , and a switching element S 2 . A connection relationship among the power supply line LH 2 , the reactor L 2 , the diode D 21 , and the switching element S 2  is the same as the connection relationship among the power supply line LH 1 , the reactor L 1 , the diode D 11 , and the switching element S 1 . In the illustration of  FIG. 2 , the switching element S 2  includes a transistor T 2  and a diode D 22 . The connection relationship between the transistor T 2  and the diode D 22  is the same as the connection relationship between the transistor T 1  and the diode D 12 . The diode D 22  is not an essential requisite, and the switching element S 2  may be, for example, a MOS field effect transistor. The conducting/non-conducting of the switching element S 2  is controlled by the control section  5 . 
     In the following, the control of the switching elements S 1 , S 2  will be described, and a primary subject thereof is the control section  5 , if not otherwise specified. 
     Here, the control section  5  is configured to include a micro computer and a storage device. The micro computer executes process steps (in other words, procedures) described in a program. The storage device mentioned above can be configured with, for example, one or a plurality of various storage devices such as a ROM (Read-Only-Memory), a RAM (Random-Access-Memory), a rewritable non-volatile memory (for example, an EPROM (Erasable-Programmable-ROM)), and a hard disk device. The storage device stores various types of information, data, and the like, and also stores the program executed by the micro computer, and also provides a work area for the execution of the program. It can be understood that the micro computer functions as various means corresponding to the process steps described in the program, or alternatively it can be understood that the micro computer implements various functions corresponding to the process steps. Here, the control section  5  is not limited to this, and various procedures executed by the control section  5 , or various means or various functions implemented by the control section  5 , may be partially or wholly implemented as hardware. 
     &lt;Single Operation of Circuits  3   a ,  3   b&gt;   
     In this switching power supply circuit, it is possible to render the switching element S 2  non-conducting so that the circuit  3   a  is singularly operated. Likewise, it is possible to render the switching element S 1  non-conducting so that the circuit  3   b  is singularly operated. Firstly, a single operation of the circuit  3   a  will be described. 
     In the circuit  3   a , when the switching element S 1  is conducting, a current flows from the input terminal P 1  to the input terminal P 2  via the reactor L 1  and the switching element S 1 . This current increases in accordance with a slope that is determined by an inductance of the reactor L 1  and the DC voltage between the input terminals P 1 , P 2  (see a current IL 1  in  FIG. 3 ). Due to this current, electromagnetic energy is accumulated in the reactor L 1 . 
     When the switching element S 1  switches from conducting to non-conducting, a current flows from the input terminal P 1  to the input terminal P 2  via the reactor L 1 , the diode D 11 , and the smoothing capacitor C 1  (see  FIG. 1 ). At this time, a voltage (induced voltage) caused by the electromagnetic energy accumulated in the reactor L 1  is added to the DC voltage between the input terminals P 1 , P 2 , and the sum thereof is applied to the smoothing capacitor C 1 . Thus, the DC voltage between the input terminals P 1 , P 2  can be boosted, and the DC voltage obtained after the boosting can be applied to the smoothing capacitor C 1 . 
     This current decreases with a slope that is based on the inductance of the reactor L 1 , an electrostatic capacitance of the smoothing capacitor C 1 , and the like (see the current IL 1  in  FIG. 3 ). Then, when this current, that is, the current IL 1 , becomes zero, the switching element S 1  is rendered conducting again. Thereafter, the above-described operation is repeated. This operation causes the current IL 1  to change while exhibiting a saw-tooth-like shape. Such a mode in which the switching element S 1  is rendered conducting immediately after the current IL 1  flowing in the reactor L 1  reaches zero is a so-called critical current mode. 
     As described above, the circuit  3   a  can function as the switching power supply circuit that boosts the voltage between the input terminals P 1 , P 2  and applies the boosted voltage between the output terminals P 3 , P 4 . Even in a time period in which a current does not flow to the smoothing capacitor C 1  (a time period in which the switching element S 1  is conducting), a current flows in the diode rectifier circuit  2 . Therefore, a conducting angle of the current flowing in the diode rectifier circuit  2  can be increased. In other words, the circuit  3   a  can function as a power factor correction circuit. 
     To achieve such switching of the circuit  3   a , the current IL 1  flowing in the reactor L 1  is detected, and the detected current IL 1  is inputted to the control section  5 . The control section  5 , for example, detects zero-crossing of the current IL 1 , and, from a time point when the zero-crossing is detected, outputs a switching signal to the switching element S 1 . Then, upon elapse of a time period determined based on an arbitrary DC voltage command value (a command value concerning the voltage between the output terminals P 3 , P 4 ), the control section  5  stops the output of the switching signal. 
     A single operation of the circuit  3   b  is similar to that of the circuit  3   a . Accordingly, the circuit  3   b  functions as a booster circuit and also functions as a power factor correction circuit. To achieve such switching of the circuit  3   b , a current IL 2  flowing in the reactor L 2  is detected, and the detected current IL 2  is inputted to the control section  5 . The control section  5 , for example, detects zero-crossing of the current IL 2 , and, from a time point when the zero-crossing is detected, outputs a switching signal to the switching element S 2 . Then, upon elapse of a time period determined based on an arbitrary DC voltage command value, the control section  5  stops the output of the switching signal. 
     Although the critical current mode is adopted in the example described above, this is not limiting. For example, it is possible that the switching element S 1  or the switching element S 2  is rendered conducting when the current IL 1  reaches a predetermined value greater than zero. This mode is a so-called continuous current mode. It is also possible that the switching element S 1  or the switching element S 2  is rendered conducting when a predetermined time period has elapsed after the time point when the current IL 1  reached zero. This mode is a so-called non-continuous current mode. Any of the modes is adoptable, and this is also true for other embodiments described later. In the following, however, the critical current mode will be described as a typical example. 
     &lt;Cooperative Operation of Circuits  3   a ,  3   b&gt;   
     In this switching power supply circuit, the circuits  3   a ,  3   b  can be cooperatively operated. This operation is also called interleaving. In the following, details will be described. 
     The switching element S 2  is rendered conducting when a predetermined time period (the predetermined time period may be zero) has elapsed after a time point when the switching element S 1  was rendered conducting. This predetermined time period is a time period shorter than a time period (hereinafter also referred to as cycle) T from when the switching element S 1  is rendered conducting to when the switching element S 1  is rendered conducting again. In the illustration of  FIG. 3 , one half of the time period T is adopted as the predetermined time period, and a case where the half of the time period T is adopted as the predetermined time period will be described below. Since a time point when the switching element S 2  is rendered conducting is determined on the basis of a time point when the switching element S 1  is rendered conducting, it can be understood that the switching element S 1  is a master-side switching element and the switching element S 2  is a slave-side switching element. 
     Due to the switching described above, the circuit  3   b  is similarly operated with a delay of one half the cycle behind the circuit  3   a . Therefore, the current IL 2  flowing in the reactor L 2  is delayed by one half the cycle relative to the current IL 1  flowing in the reactor L 1 . 
     In a case where only the circuit  3   a  is singularly operated, a current I flowing in the diode rectifier circuit  2  is equal to the current IL 1  flowing in the reactor L 1 . On the other hand, in a case where the circuits  3   a ,  3   b  are cooperatively operated, the current I flowing in the diode rectifier circuit  2  is equal to the sum of the currents IL 1 , IL 2 . In the sum, a low value portion (so-called valley) of the current IL 1  is compensated by a high value portion (so-called peak) of the current IL 2 . In the same manner, a valley of the current IL 2  is compensated by a peak of the current IL 1 . Accordingly, a variable component (so-called harmonic component) of the current I can be reduced (see the current I in  FIG. 3 ). Although a shift between the cycles of the currents IL 1 , IL 2  is not limited to one half the cycle, one half the cycle causes the greatest reduction of the harmonic component. Moreover, since the valley of the current IL 1  is compensated by the peak of the current IL 2 , the average value of the current I is increased. In other words, the maximum value of the current I can be lowered to achieve the same average value as in a case where the circuit  3   a  is singularly operated. 
     To achieve such switching of the circuits  3   a ,  3   b , the control section  5 , for example, detects zero-crossing of the current IL 1 , and, from a time point when the zero-crossing is detected, outputs a switching signal to the switching element S 1 . Then, upon elapse of a time period determined based on an arbitrary DC voltage command value, the control section  5  stops the output of the switching signal. Concurrently with this, the control section  5  outputs a switching signal to the switching element S 2  from a time point when a predetermined time period (for example, one half of the cycle) has elapsed after the time point when the zero-crossing of the current IL 1  was detected, and upon elapse of a time period determined based on an arbitrary DC voltage command value, stops the output of the switching signal. 
     As described above, in this switching power supply circuit, each of the circuits  3   a ,  3   b  can be singularly operated, and the circuits  3   a ,  3   b  can be cooperatively operated, too. The control section  5  also has a function for switching among the single operation of the circuit  3   a , the single operation of the circuit  3   b , and the cooperative operation of the circuits  3   a ,  3   b.    
     &lt;Characteristics of Circuits  3   a ,  3   b&gt;   
     In this embodiment, characteristics of at least any of the switching elements S 1 , S 2 , the diodes D 11 , D 21 , and the reactors L 1 , L 2  that belong to the circuits  3   a ,  3   b , respectively, are different from each other. Accordingly, electrical characteristics of the circuit  3   a  and electrical characteristics of the circuit  3   b  are different from each other. 
     Therefore, electrical characteristics of the switching power supply circuit in a first state where the circuit  3   a  is singularly operated, and electrical characteristics of the switching power supply circuit in a second state where the circuit  3   b  is singularly operated are different from each other. Furthermore, electrical characteristics of the switching power supply circuit in a third state where the circuits  3   a ,  3   b  are cooperatively operated is different from the electrical characteristics of the switching power supply circuit in both the first state and the second state. 
     Therefore, in this switching power supply circuit, three types of electrical characteristics can be exhibited with the two circuits  3   a ,  3   b . Moreover, it is possible to adopt a zeroth state where neither of the two circuits  3   a ,  3   b  are operated, that is, a state where both of the switching elements S 1 , S 2  are non-conducting. Adoption of this enables four types of electrical characteristics to be exhibited with the two circuits  3   a ,  3   b.    
     Hereinafter, examples of device characteristics different between the circuits  3   a ,  3   b  will be mentioned. For example, inductances of the reactors L 1 , L 2 , reverse recovery characteristics and forward voltages of the diodes D 11 , D 21 , conduction characteristics and gate constants of the switching elements S 1 , S 2 , and the like, may be mentioned. Each of these device characteristics influence any of the power factor, the efficiency, and harmonics included in the current, of the switching power supply circuit. Additionally, the influences of these device characteristics on the power factor, the efficiency, and the harmonics depend on a load (such as an output current and an output frequency) of the inverter  4 . Therefore, it is preferable, for example, to operate the switching power supply circuit while appropriately adopting the first state, the second state, or the third state in accordance with the distinction among low, medium, and high of the load of the inverter  4  (for example, the distinction of a low range, a middle range, and a high range of the output frequency of the inverter  4 ). Alternatively, it is also acceptable to adopt the zeroth state, the first state, and the third state in accordance with the distinction among low, medium, and high of the load of the inverter  4 , respectively. Needless to say, it is acceptable to adopt the zeroth to third states while distinguishing the load state of the inverter  4  into four ranges. Then, by appropriately setting the electrical characteristics of the switching power supply circuit in the zeroth state, the first state, the second state, and the third state, the switching power supply circuit can be operated in the most proper operation state in accordance with the load of the inverter  4 . In other words, this switching power supply circuit contributes to achievement of the electrical characteristics in accordance with a variation of the load. 
     As described above, the circuits  3   a ,  3   b  are illustrated as the plurality of circuits  3 . However, when N circuits  3  having different device characteristics are adopted, the number of combinations S thereof is represented by the following expression.
 
[Math. 1]
 
 S=Σ   n-I   N   N   C   n +1  (1)
 
     Here,  a C b  is represented by the following expression. 
     
       
         
           
             
               
                 
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     Accordingly, when N circuits  3  having different device characteristics are provided, S patterns of electrical characteristics of the switching power supply circuit are achieved. In other words, this switching power supply circuit contributes to achievement of the electrical characteristics in accordance with a more delicate load state. Here, some of the plurality of circuits  3  may have the same device characteristics. 
       FIG. 4  is a diagram showing one example of the switching power supply circuit. The switching power supply circuit illustrated in  FIG. 4  is different from the switching power supply circuit illustrated in  FIG. 2 , in terms of two switching elements S 1 , S 2  belonging to the circuits  3   a ,  3   b , respectively. To be specific, the switching element S 1  is an insulated gate bipolar transistor. The switching element S 2  is a MOS field effect transistor. The term “MOS” has been used for the laminated structure of metal/oxide/semiconductor in the old times, and was named from the initial letters of Metal-Oxide-Semiconductor. How ever, especially in a field effect transistor having a MOS structure, from the viewpoint of the recent improvement in the integration and the manufacturing process etc., the materials of a gate insulating film and a gate electrode have been improved. 
     For example, in the MOS field effect transistor, mainly from the viewpoint of forming the source and the drain in a self-alignment manner, polycrystalline silicon, instead of a metal, has now been adopted as the material of the gate electrode. Although a material having a high dielectric constant is adopted as the material of the gate insulating film from the viewpoint of improving the electrical characteristics, this material is not limited to oxides. 
     Therefore, adoption of the term “MOS” is not necessarily limited to a laminated structure of metal/oxide/semiconductor, and this specification does not assume such a limitation. That is, in view of common general technical knowledge, the term “MOS” herein not only is an abbreviation derived from the word origin but also has a broad sense including a stacked structure of conductor/insulator/semiconductor. 
     An insulated gate bipolar transistor has, as its device characteristics, a higher conduction loss than that of a MOS field effect transistor, and has, as its device characteristics, a larger current capacity than that of a MOS field effect transistor. 
     Accordingly, for example, in a region where the current I is low (for example, where the load of the inverter  4  is in a low range), only the circuit  3   b  is singularly operated. Since the switching element S 1  is not conducting in the region where the current I is low, occurrence of a conduction loss is suppressed and thus the efficiency is improved. For example, in a region where the current is medium (for example, where the load of the inverter  4  is in a middle range), only the circuit  3   a  is singularly operated. As a result, even if the current capacity of the switching element S 2  is deficient, damage to the switching element S 2  is avoided because the switching element S 1  having a large current capacity is used. 
     For example, in a region where the current I is high (for example, where the load of the inverter  4  is in a high range), the circuits  3   a ,  3   b  are cooperatively operated. As a result, even if the current capacity of the switching element S 1  is deficient because of the single operation of the circuit  3   a , the current is distributed to the switching elements S 1 , S 2  and therefore the current flowing in each of the switching elements S 1 , S 2  is reduced. Here, in view of a difference between the current capacities of the switching elements S 1 , S 2 , it is preferable that the maximum value of the current flowing in the switching element S 2  is lower than the maximum value of the current flowing in the switching element S 1 . This is achieved in consideration of the following two points. Firstly, when the reactor L 2  has a high inductance, the slope of the current flowing in the switching element S 2  (that is, the slope of increase of the current IL 2 ) can be reduced. Secondly, when the time period in which the switching element S 2  is conducting is short, the maximum value of the current flowing in the switching element S 2  is lowered. Accordingly, by appropriately setting the inductance of the reactor L 2  and the time period in which the switching element S 2  is conducting, the maximum value of the current flowing in the switching element S 2  can be made lower than the maximum value of the current flowing in the switching element S 1 . 
     The circuit  3   b  adopting a MOS field effect transistor as the switching element S 2  is widely used in, for example, a television receiver. Therefore, since adoption of the switching element S 2  that is a general-purpose component is allowed, the manufacturing cost can be reduced. 
     The switching element S 1  may be formed of a SiC (silicon carbide) semiconductor or a GaN (gallium nitride semiconductor), and the switching element S 2  may be formed of a Si (silicon) semiconductor. The switching element S 1  formed of a SiC semiconductor or a GaN semiconductor has a lower conduction loss than, for example, the switching element S 2  formed of a Si semiconductor. Therefore, for example, the efficiency in the first state where the circuit  3   a  is singularly operated is further enhanced. On the other hand, the manufacturing cost of the switching element S 2  formed of a S 1  semiconductor is lower than the manufacturing cost of the switching element S 1  formed by a SiC semiconductor or a GaN semiconductor. Therefore, the manufacturing cost can be reduced, as compared with a case where both of the switching elements S 1 , S 2  are formed of a SiC semiconductor or a GaN semiconductor. 
     As the reactor L 2 , a reactor having a small current capacity may be adopted. That is, the wire diameter of a coil included in the reactor L 2  may be reduced. This can downsize the reactor L 2 , and the manufacturing cost can be reduced. The circuit  3   b  adopting such a reactor L 2  is also widely used in a television receiver or the like. Therefore, since adoption of the reactor L 2  that is a general-purpose component is allowed, the manufacturing cost can be reduced. 
     &lt;Air Conditioning Apparatus mounted with Switching Power Supply Circuit&gt; 
     Next, a consideration will be given to a case where this switching power supply circuit is provided in an air conditioning apparatus. Here, similarly to the illustration of  FIG. 2 , this switching power supply circuit is provided at the input side of the inverter  4 , and the inverter  4  applies an AC voltage to a motor, not shown, to control the speed of rotation of the motor. The motor drives a compressor or a fan provided in the air conditioning apparatus. 
       FIG. 5  shows a time period during which each outside air temperature occurs in Japan, and the relationship between the outside air temperature and an air conditioning load. The air conditioning load shown in  FIG. 5  is understood as a heating load when the outside air temperature is in a range lower than 20 degrees Celsius (hereinafter, omitted), and understood as a cooling load when the outside air temperature is in a range higher than 20 degrees. As illustrated in  FIG. 5 , in a heating operation, as the outside air temperature is lower, the heating load increases, and therefore the load of the inverter  4  increases. However, as shown in  FIG. 5 , a time period during which the outside air temperature is, for example, 5 degrees or less is relatively short throughout the year. 
     Accordingly, a time period in which a high value of the heating capacity is required is short. For example, if the capacity (herein, the heating capacity) exhibited by the air conditioning apparatus when the outside air temperature is about −1 degrees (T 1 ) is defined as a rated heating capacity P 1 , an intermediate heating capacity P 2  having half the value of the rated capacity is coincident with the capacity exhibited when the outside air temperature is 7 to 8 degrees (T 2 ). A time period in which the operation is performed with the intermediate heating capacity P 2  or less (a time period in which the outside air temperature is higher than 7 to 8° C.) is longer than a time period in which the operation is performed with the intermediate heating capacity P 2  or more (a time period in which the outside air temperature is lower than 7 to 8° C.). In other words, throughout the year, the inverter  4  is frequently operated with the air conditioning load being equal to or less than the intermediate heating capacity P 2 . 
     On the other hand, in a cooling operation, as the outside air temperature is higher, a more cooling capacity is required. However, as illustrated in  FIG. 5 , a time period during which the outside air temperature is, for example, 33 degrees or higher is relatively short. Accordingly, a time period in which a high value of the cooling capacity is required is short. For example, if the capacity (herein, the cooling capacity) exhibited by the conditioning apparatus when the outside air temperature is about 35 degrees (T 4 ) is defined as a rated cooling capacity P 4 , an intermediate cooling capacity P 3  thereof is coincident with the capacity exhibited when the outside air temperature is about 29 degrees (T 3 ). A time period in which the operation is performed with the intermediate cooling capacity P 3  or less is relatively long. 
     As described above, in the air conditioning apparatus that is frequently operated with a load being medium or less, improvement in the electrical characteristics of the switching power supply circuit with respect to the load that is medium or less is particularly demanded, as compared with a case where the switching power supply circuit is adopted in other fields. 
     Therefore, this switching power supply circuit is controlled, for example, as follows. Here, a description will be given to an example case where the switching among the zeroth state, the first state, and the third state is performed in accordance with a load status. The load status is distinguished based on, for example, the current I. Accordingly, here, for example, the switching among the zeroth state, the first state, and the third state is performed in accordance with the magnitude of the current I. 
       FIG. 6  shows one example of the relationship between the magnitude of the current I and the efficiency in each state. In the illustration of  FIG. 6 , the relationships of the current I to the efficiencies in the zeroth state, the first state, and the third state are indicated by the solid curved line, the broken curved line, and the alternate long and short dash curved line, respectively. These relationships can be obtained in advance by experiments or simulations. For example, the relationship in the zeroth state can be obtained by calculating the efficiency of the switching power supply circuit while changing the current I with all the switching elements S 1 , S 2  kept non-conducting. The same is true for the first state and the third state. 
     As illustrated in  FIG. 6 , the efficiency in each state generally has an upwardly convex shape. A current value I 1  that takes the peak of the efficiency in the zeroth state is lower than a current value I 2  that takes the peak of the efficiency in the first state, and the current value I 2  is lower than a current value I 3  that takes the peak of the efficiency in the third state. Then, the efficiency in the zeroth state and the efficiency in the first state take the same value at a current value Iref 1  that is higher than the current value I 1  and lower than the current value I 2 . The efficiency in the first state and the efficiency in the third state take the same value at a current value Iref 2  that is higher than the current value I 2  and lower than the current value I 3 . 
     Accordingly, to enhance the efficiency in a wider range of currents, the switching among the zeroth state, the first state, and the third state is performed as follows. That is, when the current I is lower than the current value Iref 1 , the zeroth state is adopted by not operating the circuits  3   a ,  3   b , that is, by keeping the switching elements S 1 , S 2  non-conducting, and when the current I is higher than the current value Iref 1 , the first state is adopted by singularly operating the circuit  3   a , that is, by repeatedly switching conducting/non-conducting of the switching element S 1 . When the current I is higher than the current value Iref 2 , the third state is adopted by cooperatively operating the circuits  3   a ,  3   b , that is, by repeatedly switching conducting/non-conducting of the switching elements S 1 , S 2 . Thereby, as illustrated in  FIG. 7 , the efficiency can be enhanced in a wider range of the current I (in other words, in a wider range of the load). 
     In the illustration of  FIGS. 6 and 7 , for the purpose of comparison, the relationship between the current and the efficiency in a switching power supply circuit having the single circuit  3   a  is indicated by the alternate long and two short dashes curved line. In the illustration of  FIGS. 6 and 7 , to enhance the efficiency in the entire region of the current I, a circuit constant of the switching power supply circuit is set such that the efficiency has its peak in the vicinity of the center thereof (in the vicinity of the current value I 2 ). As understood from the illustration of  FIGS. 6 and 7 , the above-described control method can improve the efficiency in a wide range of the current I. 
     As described above, this control method can improve the efficiency in a wide range of the load. Particularly, by adopting the zeroth state when the current I is lower than the current value Iref 1 , the efficiency can be improved in a region where the load is low. Accordingly, this control method is particularly effective for an air conditioning apparatus that is frequently operated with an intermediate capacity or less. 
     When the current is coincident with the current value Iref 1 , either of the zeroth state and the first state may be adopted, and when the current is coincident with the current value Iref 2 , either of the first state and the third state may be adopted. Additionally, the criteria for switching the state are not necessarily the current values Iref 1 , Iref 2 , and a slight shift is allowed. Moreover, this control method is not limited to air conditioning apparatus, and may be adopted in a switching power supply circuit mounted on another apparatus. 
     In a case where the switching element S 1  is formed of, for example, a SiC semiconductor or a GaN semiconductor, the efficiency in the first state can be further improved as illustrated in  FIG. 8 . In the illustration of  FIG. 8 , the thin broken curved line indicates the efficiency obtained when the switching element S 1  formed of a Si semiconductor is adopted, and the thick broken curved line indicates the efficiency obtained when the switching element S 1  formed of a SiC semiconductor or a GaN semiconductor is adopted. The switching element S 1  formed of a SiC semiconductor or a GaN semiconductor also enhances the efficiency in the third state, though not shown in  FIG. 8 . 
     &lt;DC Voltage Command Value&gt; 
     As described above, the control section  5  controls the switching elements S 1 , S 2  based on the DC voltage command value that is the command value concerning the DC voltage between the output terminals P 3 , P 4  in the first to the third states. In more detail, for example, the time periods in which the switching elements S 1 , S 2  are conducting may be determined based on the DC voltage command value. 
     For example, in  FIG. 7 , the control section  5  adopts the first state when, for example, the current I is higher than the current value Iref 1 . At this time, it is preferable that the control section  5  adopts a first DC voltage command value A that is higher than the DC voltage between the output terminals P 3 , P 4  in the zeroth state. The reason therefor is as follows. That is, with the current I being higher (that is, with the load being higher), a ripple in the DC voltage increases. An increase in the ripple in the DC voltage may cause an unstable operation of the circuit  3 . To be more specific, there is a possibility that, for example, the DC voltage has a value lower than required by the inverter. In this control method, when the current I is higher than the current value Iref, the first DC voltage command value A that is still higher is adopted. This increases charges stored in the smoothing capacitor C 1 , and therefore can reduce a ripple in the DC voltage. As a result, a stable operation is achieved. On the other hand, as the DC voltage increases, a loss in the circuit  3  increases and thus the efficiency decreases. In this control method, when the current I is lower than the current value Iref, the DC voltage takes a value that is still lower. Therefore, the efficiency in this range can be improved. As described above, the efficiency of the circuit in a range where the current I is low is improved while a stable operation of the circuit  3  in a range where the current I is high is achieved. 
     In the same manner, it is desirable that a value higher than the first DC voltage command value A adopted in the first state is adopted as a second DC voltage command value B in the third state. This contributes to a stable operation of the circuit  3  in a state where the current I is higher than the current value Iref 2 . Moreover, as compared with a case where the second DC voltage command value B is adopted in a range where the current I is higher than the current value Iref 1  and lower than the current value Iref 2 , the efficiency in this range is improved. 
     Furthermore, in this embodiment, the inverter  4  drives the motor. The efficiency of the motor decreases as the current I increases. This is because an increase in the current I increases a copper loss. However, by increasing the DC voltage, the copper loss can be decreased. Although an increase in the DC voltage decreases the efficiency of the circuit  3 , the decrease in the efficiency of the motor caused thereby is smaller than the increase in the efficiency of the motor caused by decreasing the copper loss. As a result, the efficiency of the motor is improved. Accordingly, in a case where the inverter  4  drives the motor, adoption of this control method can suppress the decrease in the efficiency of the motor in a range where the current I is high, or can increase the efficiency of the motor. Additionally, as described above, this control method improves the efficiency of the circuit in a range where the current I is low, and consequently contributes to improvement in the efficiency of the motor in this range. 
     &lt;Example of Operation of Circuits  3   a ,  3   b  Having Different Device Characteristics&gt; 
     Next, a description will be given to an example of switching for cooperatively operating the circuits  3   a ,  3   b  having different device characteristics. In the following, the switching element S 1  is defined as a master-side switching element, and the switching element S 2  is defined as a slave-side switching element. 
     For example, in the switching power supply circuit of  FIG. 2 , the reactors L 1 , L 2  have different impedances. Here, it is assumed that, for example, the inductance of the reactor L 1  is twice the inductance of the reactor L 2 . Thus, the number of turns of a coil included in the reactor L 2  can be half the number of turns of a coil included in the reactor L 2 . Therefore, downsizing of the reactor L 2  is achieved, and the manufacturing cost is reduced. 
       FIG. 9  shows one example of the currents IL 1 , IL 2 , I in such a switching power supply circuit. In the circuit  3   a , in an interval between a time t 1  and a time t 3 , the switching signal is outputted to the switching element S 1  to render the switching element S 1  conducting. Accordingly, in the interval between the time t 1  and the time t 3 , the current IL 1  increases with a predetermined slope. Then, in an interval between the time t 3  and a time t 5 , the output of the switching signal to the switching element S 1  is stopped. Thereby, the switching element S 1  is rendered non-conducting, and the current IL 1  decreases with a predetermined slope. Since the current IL 1  reaches zero at a time point of the time t 5 , the switching signal is outputted to the switching element S 1  again, to render the switching element S 1  conducting. Thereafter, in the circuit  3   a , the above-described operation is repeated. 
     In  FIG. 9 , the slope of increase in the current IL 1  and the slope of decrease in the current IL 1  are, except for plus and minus, identical to each other. In other words, the current IL 1  in a case where the time period in which the switching element S 1  is conducting and the time period in which the switching element S 1  is non-conducting are identical to each other is illustrated. Therefore, in the illustration of  FIG. 9 , in one cycle T that is the time period between the time t 1  and the time t 5 , the current IL 1  has the shape of an isosceles triangle. 
     In the circuit  3   b , on the other hand, at the time t 1 , the switching signal is outputted to the switching element S 2  to render the switching element S 2  conducting. Accordingly, the current IL 2  increases with a predetermined slope. The slopes of increase in the currents IL 1 , IL 2  become smaller as the inductances of the reactors L 1 , L 2  are higher. Here, the inductance of the reactor L 2  is half the value of the inductance of the reactor L 1 , and therefore the slope of increase in the current IL 2  (the ratio of increment thereof relative to time) is twice the slope of increase in the current IL 1 . 
     At a time t 2  at which the maximum value of the current IL 2  and the maximum value of the current IL 1  are coincident with each other, the output of the switching signal to the switching element S 2  is stopped. Thereby, the switching element S 2  is rendered non-conducting, and the current IL 2  decreases with a predetermined slope. The slopes of decreases in the currents IL 1 , IL 2  also become smaller as the inductances of the reactors L 1 , L 2  are higher, respectively. Here, the inductance of the reactor L 2  is half the value of the inductance of the reactor L 1 , and therefore the slope of decrease in the current IL 2  (the ratio of decrement thereof relative to time) is twice the slope of decrease in the current IL 1 . 
     In view of the relationships of the slopes of the currents IL 1 , IL 2 , the current IL 2  reaches zero at a time point (that is, the time t 3 ) when the half of one cycle T elapses. Then, at the time t 3 , the switching signal is outputted to the switching element S 2  again, and the above-described operation is repeated. 
     Due to this cooperative operation of the circuits  3   a ,  3   b , the current I (=IL 1 +IL 2 ) flows in the diode rectifier circuit  2 . 
     Since the currents IL 1 , IL 2  have their valleys at the same time (for example, at the times t 1 , t 5 ), the current I that is the sum of them also has its valley at this time. In other words, at this time, the minimum value of the current I is zero that is equal to the minimum values of the currents IL 1 , IL 2 . Additionally, at the times t 2 , t 4 , the maximum value of the current I takes a value higher than the maximum values of the currents IL 1 , IL 2  (here, 1.5 times higher than those of the currents IL 1 , IL 2 ). Thus, there is a large difference (here, 1.5 times the currents IL 1 , IL 2 ) between the maximum value and the minimum value of the current I that is the sum of the currents IL 1 , IL 2 . Therefore, harmonics are likely to occur. 
     It may be also acceptable to shift the time points when the switching elements S 1 , S 2  are rendered conducting. In other words, it may be acceptable to change a phase difference between the currents IL 1 , IL 2 . However, even if the phase difference is changed, the difference between the maximum value and the minimum value of the current I is not changed so much. 
     Therefore, it may be possible that, as shown in  FIG. 10 , the switching signal is outputted to the switching element S 2  only in a time period from the time t 4  to the time t 5  during one cycle T from the time t 1  to the time t 5 . This causes the current IL 2  to make a peak in a time period in which the current IL 1  makes a valley, and the current IL 2  to be zero in a time period in which the current IL 1  makes a peak. Thus, the difference between the maximum value and the minimum value of the current I that is the sum of the currents IL 1 , IL 2  can be reduced. Here, the maximum value and the minimum value of the current I are equal to and 0.5 times the maximum value of the current IL 1  (or the current IL 2 ), respectively, and the difference therebetween is reduced to 0.5 times the maximum value of the current IL 1  (or the current IL 2 ). 
     In a case where the ratio of the inductances of the reactors L 1 , L 2  is 2, as illustrated in  FIG. 10 , the switching element S 2  is rendered conducting only in the last one-fourth of the one cycle T. In a case where the ratio of the inductance of the reactor L 1  relative to the inductance of the reactor L 2  is N (N is a number greater than 1), the switching element S 2  may be rendered conducting only in the last one 2N-th of the one cycle T. In other words, the switching element S 2  may be rendered conducting only in an interval between a third time point (t 4 ) that is prior, by a time period corresponding to one N-th of the one cycle, to a second time point (t 5 ), and the second time point (t 5 ) that is the next time after the first time point (t 1 ) to render the switching element S 1  conducting. 
     For example, in the switching power supply circuit of  FIG. 2 , the reactors L 1 , L 2  may have different inductances and different current capacities. Here, for example, it is assumed that the inductance of the reactor L 1  is lower than the inductance of the reactor L 2 , and the current capacity of the reactor L 1  is smaller than the current capacity of the reactor L 2 . This can make the wire diameter of a coil included in the reactor L 2  smaller than the wire diameter of a coil included in the reactor L 2 . Accordingly, even though the reactor L 2  has a high inductance, an increase in the size of the reactor L 2  is suppressed, and additionally the manufacturing cost is reduced. 
       FIG. 11  shows one example of the currents IL 1 , IL 2 , I in such a switching power supply circuit. The switching of the switching element S 1  of the circuit  3   a  is the same as the switching having been described with reference to  FIG. 9 . 
     On the other hand, to the switching element S 2 , the switching signal is outputted in a time period (that is, one half the cycle) from the time t 3 , that is when one half the cycle elapses after the time t 1  at which the switching element S 1  has been rendered conducting, to the time t 5 . In this manner, the switching element S 2  is rendered conducting. Therefore, in the time period from the time t 3  to the time t 5 , the current IL 2  increases. However, since the inductance of the reactor L 2  is higher than the inductance of the reactor L 1 , the slope of the current IL 2  is smaller than the slope of the current IL 1 . Since the time period in which the switching element S 1  is rendered conducting and the time period in which the switching element S 2  is rendered conducting are the same, the maximum value of the current IL 2  is lower than the maximum value of the current IL 1 . 
     Thus, even if the current capacity of the reactor L 2  is smaller than the current capacity of the reactor L 1 , the circuits  3   a ,  3   b  can be cooperatively operated without causing a malfunction of the reactor L 2 . Needless to say, the current capacity of the reactor L 2  is set larger than the current of the current IL 2 . 
     Here, also in the illustration of  FIG. 11 , it may be acceptable to shift the time points when the switching elements S 1 , S 2  are rendered conducting. In other words, it may be acceptable to change the phase difference between the currents IL 1 , IL 2 . 
     For example, in the switching power supply circuit having been described with reference to  FIG. 9 , it may be possible to make the inductance of the reactor L 2  closer to the inductance of the reactor L 1 . This reduces the slope of the current IL 2 , and thus reduces the maximum value of the current IL 2 . Such currents IL 1 , IL 2 , I are illustrated in  FIG. 12 . Along with the reduction in the maximum value of the current IL 2 , the maximum value of the current I is also reduced. Thereby, the difference between the maximum value and the minimum value of the current I is reduced, and additionally the current capacity of the reactor L 2  is reduced. 
     Also in the illustration of  FIG. 12 , it may be acceptable to shift the time points when the switching elements S 1 , S 2  are rendered conducting. In other words, it may be acceptable to change the phase difference between the currents IL 1 , IL 2 . 
     While the invention has been described in detail, the foregoing description is in all aspects illustrative and not restrictive. It is therefore understood that numerous modifications and variations, though not illustrated herein, can be devised without departing from the scope of the invention. 
     DESCRIPTION OF THE REFERENCE NUMERALS 
     
         
         
           
             D 11 , D 12  diode 
             L 1 , L 2  reactor 
             LH 1 , LH 2 , LL power supply line 
             P 1 , P 2  input terminal 
             P 3 , P 4  output terminal 
             S 1 , S 2  switching element