Patent Publication Number: US-10770975-B2

Title: Resonant power converters with switchable resonant modes

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 15/095,021, filed on Apr. 9, 2016, which is a continuation-in-part of U.S. patent application Ser. No. 14/185,885 filed on Feb. 20, 2014, now U.S. Pat. No. 9,344,042B2, which is related and claims priority to U.S. Provisional Application No. 61/850,972, titled, “High Efficiency Power Amplifiers with Advanced Power Solutions” filed on Feb. 27, 2013, which is herein incorporated by reference. 
    
    
     TECHNICAL FIELD 
     The present invention relates to power conversion and power amplifiers, and, in particular embodiments, to high efficiency power topologies and control suitable for high efficiency power amplifiers and power converters. 
     BACKGROUND 
     Power amplifiers are widely used in wireless communication systems and other electronic devices, especially in mobile devices. To achieve high system efficiency and/or longer battery life, it is very important to maintain high efficiency in power amplifiers. 
     The power amplifiers in many systems see signals with high peak to average power ratio. Because a typical power amplifier gets its energy from a power supply coupled to the drain (or the collector if a transistor is used) of its main power switch (a MOSFET or a transistor), its power efficiency can be improved by changing the drain voltage of its main power switch through changing the voltage of its drain power supply according to the envelope of the signal being processed by the power amplifier. When the signal envelope has high bandwidth as often seen in today&#39;s wireless systems, changing the voltage of drain power supply may cause the output signal of the power amplifier to be distorted. 
     Moreover, as the bandwidth of the signal increases, the power supply&#39;s control bandwidth should be increased accordingly. This put very high burden into the power supply, and the efficiency of the power supply is low with existing technology. 
     Improvements are needed to reduce the signal distortion in a power amplifier with variable drain voltage, and to increase the efficiency of the drain power supply. 
     SUMMARY OF THE INVENTION 
     These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by preferred embodiments of the present invention which provides an improved resonant power conversion. 
     According to one embodiment of this disclosure, a system includes an input port having an input voltage, an output port having an output voltage, and a power converter having a switch network with a plurality of power switches and a first resonant tank having a first resonant capacitor and a first resonant inductor, where at least one resonant component within the first resonant capacitor and the first resonant inductor is a switchable component configured to switch between different values. The system further includes a resonant mode selection block configured to adjust a value of the switchable component to maintain a performance of the system, and a controller configured to adjust a switching frequency or a duty cycle of the power converter. 
     According to another embodiment of this disclosure, an apparatus includes an input port having an input voltage, an output port having an output voltage, and a transformer having a primary winding and a secondary wingding, where the primary wingding is coupled to the input port. The apparatus further includes a power switch coupled between the primary winding and the input port, a rectifier circuit coupled between the secondary winding and the output port, a primary resonant tank having a primary resonant capacitor in parallel with the power switch and a primary resonant inductor including the primary winding, and a secondary resonant tank having a secondary resonant capacitor coupled between the secondary winding and the rectifier circuit, and a secondary resonant inductor comprising the secondary winding, The apparatus also includes a control block configured to control at least one of a duty cycle and a switching frequency of the power switch. 
     According to yet another embodiment of this disclosure, a method includes configuring a power converter comprising an input port having an input voltage, an output port having an output voltage, a switch network having a plurality of power switches, and a resonant tank comprising a resonant capacitor and a resonant inductor, where at least one resonant component within the resonant capacitor and the resonant inductor is a switchable component configured to switch between different values. The method further includes configuring a resonant mode selection block to adjust a value of the switchable component to maintain a performance of the system, and configuring a controller to adjust a switching frequency or a duty cycle of the power converter. 
     The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures or processes for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  illustrates a schematic diagram of a power amplifier; 
         FIG. 2  illustrates a schematic diagram of a first illustrative embodiment of a power amplifier in accordance with various embodiments of the present disclosure; 
         FIG. 3  illustrates a detailed schematic diagram of an embodiment of a power converter in accordance with various embodiments of the present disclosure; 
         FIG. 4  illustrates a schematic diagram of an embodiment of a power amplifier in accordance with various embodiments of the present disclosure; 
         FIG. 5  illustrates an embodiment of an envelope reference in accordance with various embodiments of the present disclosure; 
         FIG. 6  illustrates an embodiment of a power amplifier in accordance with various embodiments of the present disclosure; 
         FIG. 7  illustrates an embodiment of a power amplifier with various embodiments of the present disclosure; 
         FIG. 8  illustrates simulated waveforms of a power amplifier in accordance with various embodiments of the present disclosure; 
         FIG. 9  illustrates an embodiment of a power supply architecture in accordance with various embodiments of the present disclosure; 
         FIG. 10  illustrates embodiments of basic cells of an asymmetric multi-level power synthesizer in accordance with various embodiments of the present disclosure; 
         FIG. 11( a )  illustrates embodiments of an asymmetric multi-level power synthesizer in accordance with various embodiments of the present disclosure; 
         FIG. 11( b )  illustrates embodiments of an asymmetric multi-level power synthesizer in accordance with various embodiments of the present disclosure; 
         FIG. 12  illustrates an embodiment of a multi-level power synthesizer with linear control in accordance with various embodiments of the present disclosure; 
         FIG. 13  illustrates an embodiment of a multi-output power converter in accordance with various embodiments of the present disclosure; 
         FIG. 14( a )  illustrates an embodiment of a resonant power converter with automatic frequency tracking in accordance with various embodiments of the present disclosure; 
         FIG. 14( b )  illustrates simulated waveforms of the resonant power converter in  FIG. 14( a )  in accordance with various embodiments of the present disclosure; 
         FIG. 15( a )  illustrates an embodiment of a resonant gate drive in accordance with various embodiments of the present disclosure; 
         FIG. 15( b )  illustrates the state-plane trajectory of the resonant gate drive in  FIG. 15( a )  in accordance with various embodiments of the present disclosure; 
         FIG. 15( c )  illustrates an embodiment of a resonant gate drive with gate voltage shifting in accordance with various embodiments of the present disclosure; 
         FIG. 15( d )  illustrates simulated waveforms of the resonant gate drive in  FIG. 15( c )  in accordance with various embodiments of the present disclosure; 
         FIG. 16  illustrates an embodiment of a resonant power converter with outputs arranged in pairs in accordance with various embodiments of the present disclosure; 
         FIG. 17( a )  illustrates an embodiment of a current-fed power converter in accordance with various embodiments of the present disclosure; 
         FIG. 17( b )  illustrates simulated waveforms of the power converter in  FIG. 17( a )  in accordance with various embodiments of the present disclosure; 
         FIG. 18  illustrates an embodiment of a multi-output power supply with voltage regulation in accordance with various embodiments of the present disclosure; 
         FIG. 19  illustrates an embodiment of a multi-output current-fed full bridge power converter in accordance with various embodiments of the present disclosure; 
         FIG. 20  illustrates simulated waveforms of the power converter in  FIG. 19  in accordance with various embodiments of the present disclosure; 
         FIG. 21( a )  illustrates embodiments of a multi-output power converter based on a buck topology in accordance with various embodiments of the present disclosure; 
         FIG. 21( b )  illustrates embodiments of a multi-output power converter based on a buck topology in accordance with various embodiments of the present disclosure; 
         FIG. 22  illustrates an embodiment of a dynamic power supply based on a buck topology in accordance with various embodiments of the present disclosure; 
         FIG. 23  illustrates an embodiment of a block diagram of a resonant power supply in accordance with various embodiments of the present disclosure; 
         FIG. 24  illustrates an embodiment of a half-bridge resonant power converter in accordance with various embodiments of the present disclosure; 
         FIG. 25  illustrates an embodiment of a flyback resonant power converter in accordance with various embodiments of the present disclosure; 
         FIG. 26  illustrates an embodiment of a half-bridge resonant power converter in accordance with various embodiments of the present disclosure; 
         FIG. 27  illustrates an embodiment of a gate drive circuit for a power converter in accordance with various embodiments of the present disclosure; and 
         FIG. 28  illustrates an embodiment of a phase diagram of the gate drive circuit shown in  FIG. 27  in accordance with various embodiments of the present disclosure. 
     
    
    
     Corresponding numerals and symbols in the different figures generally refer to corresponding parts unless otherwise indicated. The figures are drawn to clearly illustrate the relevant aspects of the various embodiments and are not necessarily drawn to scale. 
     DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS 
     The making and using of the presently preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention. 
     The present invention will be described with respect to preferred embodiments in a specific context, namely in power amplifiers and power supply technologies for power amplifiers. The invention may also be applied, however, to a variety of other electronics systems, including integrated circuits, CPUs (central processing units), computers, telecom equipments, any combinations thereof and/or the like. Hereinafter, various embodiments will be explained in detail with reference to the accompanying drawings. 
     Often a power amplifier is required to process a signal with a variable amplitude envelope which is related to the information to be communicated. The envelope may have high-frequency components up to several tens or even hundreds of MHz in radio frequency (RF) communication systems and devices and other equipment. The carrier frequency is usually much higher than the highest component frequency of the envelope. In the last few decades, people have tried to develop high efficiency technologies, such as envelope tracking (ET) and envelope elimination and restoration (EER), for power amplifiers intended for variable envelope applications.  FIG. 1  shows a block diagram for such a high efficiency amplifier. S 1  is the main switch. Although a MOSFET is shown as an example, it can also be a bipolar transistor (BJT) or other switching devices. L 1  is the RF choke, C 2  is the DC blocking capacitor to prevent dc component from getting into the RF output. C 1  is the bypass capacitor to filter out noise in the power path. The envelop power supply  101  provides drain power to the switch, and is designed to follow the envelope information of the signal to be amplified. The bias circuit sets up the switch&#39;s operating point, and RF signal circuit conditions the signal to be amplified and couples it to the switch. The impedance match circuit  104  matches the impedance of S 1  with the impedance of the RF load connected to the power amplifier. 
     With EER technology (such as in polar modulation), S 1  works in switching mode (such as in Class E mode). The envelope power supply&#39;s output Vdd is basically a replica of the envelope of the RF signal, and the RF signal circuit for the gate (or base if a BJT is used) in the power amplifier just processes the phase information of the RF input. If the timing of the envelop signal is aligned perfectly with the phase signal, the RF output can have the right envelope and phase information. Because in the switching mode the switch S 1  works in two low loss states: saturation (ON) or block (OFF), the EER technique theoretically has high power efficiency. However, it is extremely difficult to achieve a good phase and envelope alignment, and it&#39;s very challenging to implement a fast envelope power supply, so it&#39;s very expensive and difficult to implement EER technology in practice. 
     With ET technology, S 1  works in linear mode (such as in Class A, Class AB or Class B mode). Vdd tracks but is slightly higher than the amplitude of RF output signal, and the gate signal is the same as in a normal linear amplifier (i.e. has both amplitude and phase information). With ET technology, the transistor&#39;s power loss is greatly reduced compared to that in a normal linear power amplifier due to the reduced voltage applied to the transistor. As a result, the ET power amplifier also achieves high power efficiency. ET is a very promising technology for high efficiency linear power amplifiers, but similar to the limitations of EER technology mentioned above, ET technology also faces tremendous challenges: 
     The envelope power supply has to provide power with very wide frequency spectrum. For example, in LTE technologies, the RF signal spectrum may have a bandwidth up to 20 MHz, but the envelope signal spectrum may have an effective bandwidth up to 100 MHz. The state of art power technology cannot process such wide-bandwidth power effectively; 
     The fast changing envelope power applied to the drain path of an ET power amplifier causes significant RF signal distortion, and thus tremendous predistortion needs to be used to regain the linearity of the amplification. Such predistortion normally requires a large amount of calibration work and complex procedure, and thus significantly increases the implementation cost. 
     The current ET technology doesn&#39;t consider the internal dynamics of a power amplifier to the drain power, and thus cannot be optimized for efficiency improvement reliably. 
     In this disclosure, new technologies addressing the above issues will be proposed and discussed in details. Although the description is made in the context of ET technology, the principles can also be applied to EER technology and polar modulation, as they also involve interaction between the fast envelope power supply and the power amplifier it powers, and the need of very fast dynamic power supplies. 
       FIG. 2  shows a simplified block diagram of a new ET+ technology. Although a single transistor Class A power amplifier is shown, similar technologies can be applied to other power amplifier topologies, such as Class AB, Class B, Push-pull, Class E, Class F etc. Compared to the topology in  FIG. 1 , the ET+ topology has an envelope interface circuit  105  between the power switch and the optional impedance matching circuit  104 . In an embodiment in accordance with the present invention, the envelope interface circuit  101  can be part of the impedance matching circuit  104 . As well known in the industry, harmonic termination function can be also part of the impedance matching circuit. The envelope interface circuit consists of an envelope frequency shunt  106  and a carrier frequency path circuit  107 . The circuit  106  presents a low impedance path to the envelope signal, thus allowing the envelope current to flow through it and return to the envelope power supply  101  with little impact to the output RF signal. The circuit  106  presents a high frequency to the carrier frequency, and thus doesn&#39;t impact RF signal amplification significantly. On the contrary, the carrier frequency path circuit  107  presents a high impedance path to the envelope signal, forcing the envelope signal to go through circuit  106 . To carrier frequency signal, the circuit  107  presents a low impedance path, allowing the RF output signal to pass through it without much distortion or attenuation. With this architecture, clear envelope signal path (with relatively lower frequency contents) and carrier frequency path (with relatively higher frequency contents) are established. As a result, the variable amplitude power applied to the drain path of the power switch may not cause significant distortion to the RF output, significantly reducing the requirements of predistortion and its associated costs. This concept can be easily extended to other power amplifier architectures, such as EER and polar modulation. 
       FIG. 3  shows a more detailed block diagram of the ET+ technology. The envelope interface circuit  106  is implemented with simple circuits. In one embodiment, the envelope frequency shunt circuit  106  is implemented as an inductor L 2  and an optional resistor R 1 . R 1  also damps possible resonance between reactive components, and so improves the signal quality in the amplifier. Moreover, R 1  allows the drain voltage to change relatively quickly with the change of the bias current or bias voltage to the switch S 1  when needed, and thus improves the envelope controllability of the architecture. 
     One of the key aspects to achieve high performance and high efficiency in power amplifiers is to operate the switch at the right operating point. As a minimum requirement, the valleys (the lowest points of the waveform) of the drain voltage should be just slightly higher than zero at any point of time, and the valleys of the source current should be just slightly higher than zero under all conditions too. This requires the switch be biased according to the envelope of the RF input (or output) signal. As the envelope bandwidth increases over time for higher rates of information transfer, the drain voltage envelope of the switch (Vd) can have significant deviation from the power source voltage Vdd due to the complex dynamics of the power amplifier. As the envelope bandwidth is much lower than the carrier frequency, the envelope dynamics in  FIGS. 3 and 4  is determined mainly by L 1 , C 2 , L 2  and R 1 , and also depends on the characteristics of the power switch S 1 . Proper control is needed to ensure the operating point of S 1  to follow the signal envelope properly. In  FIG. 4 , an envelope reference block  111  converts the RF signal amplitude at input or output of the power amplifier to envelope reference signals Vdr for the drain voltage and Idr for the source current of S 1 . The reference is based on the signal envelope, and a buffer signal is added to the signal envelope to make sure that the envelope reference is higher than the required signal envelope by a certain margin, as is exemplified by  FIG. 5 . The buffer value can be fixed or adjusted according to actual operation so that the performance can be optimized. Also, the reference signals may not have the exact replica of the envelope signal, and some of the highest frequency components in the envelope signal may be reduced or removed from the reference signals to reduce implementation complexity and cost. An envelope detect circuit  113  is coupled to the drain of S 1 , so its voltage envelope Vd is sensed and used as a feedback signal. Similarly, the source current of S 1  is sensed and coupled to an envelope detect circuit  114  to generate the envelope signal Is of the source current. A control block  112  may consist of feed-forward function, feedback function and any combination thereof. The block  112  processes the reference signals Vdr, Isr and feedback signals Vd and Is, to generate the drain power reference Vddr, and bias control reference Vc. The dynamic power supply  101  generates a drain power Vdd according to Vddr, while the bias control circuit  102  uses Isr as the reference to control the bias of S 1 . The RF signal conditioning and control, which may involve functions such as amplification, delay, filtering, alignment etc are performed by the RF signal circuit and are not discussed in details here, as they do not directly contribute to the envelope control. Although voltage signals are used to describe the envelope signal Vd, Is and their references, some or all of these signals can also be current signals. 
     The control architecture in  FIG. 4  can have many different implementations.  FIG. 6  shows a preferred embodiment, in which both the drain voltage envelope and source current envelope are controlled through feedback control. A difference block generates the drain voltage envelope error signal by comparing the reference signal Vdr and the envelope feedback signal Vd. Then the error signal is processed by compensation block  115  to generate the drain power reference Vddr. Another difference block generates the source current envelope error signal by comparing the reference signal Isr and the envelope feedback signal Is. Then this error signal is processed by compensation block  117  to generate the bias control voltage Vc.  FIG. 7  shows another preferred embodiment in which both feedback control and feed-forward control are used to accomplish the envelope control function. In accordance with this embodiment, a feed-forward control block  118  is used to process the reference signal Vdr. Such feed-forward function can consider known effects of the power amplifier dynamics. However, due to parameter variations and other deviation from ideal condition, it may not be practical and economical to rely on the feed-forward control alone for the drain voltage envelope control. Therefore, a feedback control block  116  similar to the one in  FIG. 6  is also included to fine tune the control action for the drain voltage. The outputs of the block  118  and the block  116  are summed together to adjust the output of the dynamic power supply  101 . Because of the added feed-forward control, the operating range and requirements of the feedback control in  FIG. 7  can be lower than that in  FIG. 6 . Since the low-frequency dynamics between the source current and gate voltage of a switch is usually simpler than the relationship between the drain voltage and power input, it&#39;s possible to use just feed-forward control for the source current envelope control, as is shown in  FIG. 7  (through feedforward block  119 ). However, if the effect of the drain voltage on the gate voltage (the Miller effect) is to be considered, an optional link can be established by sending the drain envelope signal Vd into the feed-forward block for the source current control, so proper measures can be taken to reduce or eliminate the miller effect in the source current. 
       FIG. 8  shows simulated waveforms of a drain voltage envelope and its reference in an embodiment. Clearly, with proper design the drain voltage envelope can be controlled to follow its reference closely. 
     A key aspect of the above implementation is to implement a high-efficiency dynamic power supply with very wide bandwidth. Generally, to achieve a high efficiency power conversion, a switching power converter is preferred over a linear power supply. However, a state of art switching power converter may not be able to have wide enough bandwidth. Some people proposed to use a combination of switching power converter and a linear power supply in a series or parallel configuration. Such combination naturally has relatively low efficiency and high control complexity. Another proposed method is to use a multi-output power supply followed by a voltage selection circuit, and the voltage selection circuit selects one of the voltages as the output of the dynamic power supply. However, to achieve a good result, the number of the voltages should be high, thus resulting in high implementation cost. 
     A new power architecture is shown in  FIG. 9  for fast dynamic power supplies. A multi-output power supply  120  with a limited number of different output voltage levels are followed by an asymmetric multi-level power synthesizer  130 . The power synthesizer  130  adds its input voltages in different ways so that its output voltage has a much higher number of voltage levels than its number of input voltages. 
       FIG. 10  shows the basic cells of an asymmetric multi-level synthesizer.  FIG. 10( a )  is a two-level cell. Its output is zero when S 1  is ON, and V 1  when S 2  is ON. S 1  and S 2  are controlled to be in complementary mode with a short transition between states in which both switches are off to avoid cross conduction.  FIG. 10( b )  shows a four-level cell, with an asymmetric input voltage pair  152 , and a four-switch network  151  consisting of a switch pair S 1  and S 2  in a totem pole configuration, and a second switch pair S 3  and S 4  in another totem pole configuration. An output terminal is coupled to the common node of each totem pole (i.e. at the junction of the top switch and the bottom switch in the pair). Two voltage sources at the input can have a substantially doubling relationship as described above: V 2 =2*V 1 . There are 4 possible states: 
     S 1  and S 4  OFF, S 2  and S 3  ON: Vo=0; 
     S 1  and S 3  ON, S 2  and S 4  OFF: Vo=V 1 ; 
     S 2  and S 4  ON, S 1  and S 3  OFF: Vo=V 2 =2V 1 ; 
     S 1  and S 4  ON, S 2  and S 3  OFF: Vo=V 1 +V 2 =3V 1   
     Again, the S 1  and S 2  switch pair, as well as the S 3  and S 4  switch pair, are controlled to be in complementary mode so at most one switch conducts in a pair at any time. There is a short transition period if the switches in a pair need to change states during which both switches are off to avoid cross conduction. With this structure, if the input voltages of synthesizer have a doubling relationship between them (such as V 2 =2*V 1 , V 3 =2*V 2  and so on), the synthesizer basically becomes a binary encoder of voltages (power), and the levels of the output voltage have even step size between them. For example, if the multiple-output power supply&#39;s output includes 1V and 2V, then the output of the synthesizer can be 0V, 1V, 2V, and 3V. Generally, n voltages at the input of the synthesizer can be synthesized into 2 N  different voltage levels at the output using 2N switches. However, in other embodiments the input voltages may not follow the doubling relationship precisely and the step size between adjacent voltage levels does not have to be the same in the whole range. Generally, it is desired that the doubling relationship substantially to achieve good performance. For example, the higher voltage in a four-level cell should be maintained within a range of 1.5 to 2.5 times the lower voltage. Although the doubling relationship will be used in the following description, the principles of power conversion, power synthesizer and control can be easily modified for other voltage relationships. 
     Any number of two-level cells and four-level cells can be connected in series by putting their output voltages in series to get a higher number of voltage levels. For example, a configuration with a two-level cell in series with a four-level cell makes an 8-level synthesizer, and a configuration with a four-level cell directly in series with another four-level cell makes a 16-level synthesizer. Note that the input voltages of the adjacent cells should also follow a voltage doubling relationship, i.e. the lower input voltage in the cell with higher voltages should be twice the higher input voltage in the cell with lower voltages. More voltage levels can be obtained by putting more cells in series, as this structure is easily expandable. Generally, a switch network with a N1-switch cell and a N2-switch cell is capable of producing N1 times N2 voltage levels. When one more cell with N3 switches is added, the number of voltage levels the switch network can produce is increased to N3 times the original number of voltage levels. Also, if a switch network has N input voltages, it is capable of producing 2 N  voltage levels using 2N switches.  FIG. 11( a )  shows examples of an 8-level synthesizer, and  FIG. 11( b )  shows a 16-level synthesizer. The impedance network  141  is coupled between the switch networks and the output and reduces the current flowing through the switches during the switching transition between different states, and thus increases the efficiency. The impedance network has naturally filtering function working together with the output capacitor Co. In one preferred embodiment, the impedance network is an inductor, names as L 1 . L 1  and the output capacitor Co may become part of a resonant tank. The value of L 1  should not be too high to avoid significant voltage oscillation at the output. In a preferred embodiment, the characteristic impedance 
               Zo   =           L   ⁢           ⁢   1     Co       ≈   Rs       ,         
where Rs is the resistance in the current path inside the power synthesizer, including the switch resistances, connection resistance, and resistance of L 1 . This means the resonant tank consisting of L 1  and C 1 , as well as Rs has a quality factor (defined as Zo/Rs) of substantially close to 1, for example in the range of 0.4 to 2.5. L 1  can be a discrete inductor, or the parasitic inductance in the current path of the power synthesizer including the package inductances of the switches, and any combination thereof. In a preferred embodiment, Co can be placed at or inside the power amplifier the synthesizer powers. If the synthesizer and the power amplifier are in different dies or different packages, the parasitic inductance from the packages and interconnection between them may be enough to serve the purpose of L 1 .
 
     As explained above, an asymmetric power synthesizer behaves like a power encoder. The drive signals for the switches in a power synthesizer can be obtained from a decoding process of the digital representative of the desired voltage. If the desired voltage is not in a digital format originally, an ADC converter can be used to digitize the required value. This control scheme is especially suitable for digital implementation. 
     The output voltage of a power synthesizer as described above has discrete values as it changes from one level to another level rapidly. Sometimes it&#39;s desirable to have a smoother and more accurate output. A linear regulator can be added at the output to achieve the smoothing function during a level change and/or to realize accurate voltage control. This linear regulation can also be implemented by controlling the gate drive voltage amplitude of switches within the synthesizer.  FIG. 12  shows an example, with a four-level symmetric cell  150  and another four-level cell  151 , and an impedance cell  141 , and output voltage C 0 . Again, the impedance network  141  reduces the current flowing through the switches during the switching transition between different states, and thus increases the efficiency. The impedance network has naturally filtering function working together with the output capacitor Co. In one preferred embodiment, the impedance network is an inductor, names as L 1 . L 1  and the output capacitor Co may become part of a resonant tank. A control circuit  142  is used to adjust the amplitude of the driving voltage of S 1  and S 2 . The timing of S 1  and S 2 &#39;s gate drive is still controlled by the decoder discussed above, and other switches are still controlled by the decoder. The linear control of S 1 &#39;s and S 2 &#39;s drive voltage changes the resistance of this switch pair, and thus can achieve fine-tune regulation of the output voltage of the power synthesizer. This linear control can also be used to reduce the output oscillation if needed. Obviously, the range of the linear adjustment of the output voltage can be less than 1 step size of the voltage levels, and thus high efficiency can still be achieved. This is a simple and low cost implementation of accurate and fast dynamic control in dynamic power supplies. 
     The power synthesizer needs multi voltage rails at the input. These voltage rails can be generated by multi-output power supplies. Such power supplies can be implemented with different topologies, such as flyback converters, forward converters, etc.  FIG. 13  shows an arrangement with a flyback topology with 2 pairs of output voltages ( 171  and  172 , more outputs can be added easily by repeating the pattern). If the relationship between the number of turns of the secondary windings has a doubling relationship:
 
 n 2=2* n 1, n 3=2* n 2, n 4=2* n 3
 
Then ideally the output voltage will also have a doubling relationship:
 
 V 2=2* V 1, V 3=2* V 2, V 4=2* V 3
 
     By controlling of duty cycle of the primary switch Sp 1 , the output voltages can be regulated to the desired value. S 1  through S 4  in the secondary are synchronous rectifiers and can be replaced with diodes if desired. The output voltages can be easily arranged into asymmetric voltage pairs ( 171  and  172 ) to suit the needs of a power synthesizer. 
     In some applications, the input voltage Vin may already be at the right value for the system, so the multi-output power supply just needs to convert the input voltage to needed multi-output format, and does not need to adjust values of its outputs. In this case the multi-output power supply behaves as a multi-output bus converter.  FIG. 14( a )  shows a multi-output resonant converter with local compensation suitable for such applications. Half-bridge configurations at the primary side ( 180 ) and at the secondary side ( 181  and  182 ) are shown in the figure. However, as well known in the industry, full-bridge and other suitable configurations, and any combination thereof, can also be used. Traditionally, a resonant converter such as an LLC converter has a resonant tank only at the primary side of the transformer. When such a topology is extended to multi-output configuration by adding more transformer windings, each output&#39;s voltage is affected by the leakage inductance of the secondary winding to which it is coupled. Therefore, the voltage of the each output may deviate significantly from the ideal value determined by the turns-ratio of the transformer windings, resulting in voltage inaccuracy. In the new topology of  FIG. 14 , each output also has a resonant tank coupled to the associated secondary winding, consisting of two resonant capacitors in half-bridge configuration (e.g. Cr 11  and Cr 12  for V 1 ) and a resonant inductor (e.g. Lr 1  for V 1 ). This double-side resonance topology significantly reduces the effect of transformer leakage inductance on voltage regulation, and thus achieves better cross regulation between different outputs, and also allows the output voltage to be sensed at the primary side more accurately (from feedback winding  161 , for example). The resonant inductors can be discrete parts, or the leakage inductance of the transformer windings, or any combination thereof. Series resonant tanks are shown in the figure, but other types of resonant tanks can also be used, including LLC (utilizing the magnetizing inductance of the transformer), LCC, parallel resonant, series-parallel resonant, multi-element resonant etc. However, the resonant tanks at the primary side and the secondary side should have the same configuration, and the components in the resonant tanks should have about the same equivalent values (considering the scaling factor of transformer&#39;s turns ratios). All the resonant tanks should be designed to have the same or approximately the same resonant frequency (or resonant frequencies if more than one resonant frequencies existing in a tank). If the converter&#39;s switching frequency is the same as the designed resonant frequency, and thus the leakage inductance (or any added inductance to the resonant tanks) is compensated by the associated resonant capacitors, and resonant tanks have a voltage gain of 1. Under this condition, the output voltages follow the exact relationship determined by the transformer winding turns-ratios, if the power losses of the components are ignored: 
                 V   ⁢           ⁢   2     =         n   ⁢           ⁢   2       n   ⁢           ⁢   1       ⁢   V   ⁢           ⁢   1       ,         
where n 2  and n 1  are the number of turns of the secondary windings coupled to V 2  and V 1  respectively.
 
     Even if the operation frequency is not exactly the same as the resonant frequency of each resonant tank due to component tolerance and other issues, the compensation effect of the resonant capacitors can still improve the performance of the power supply. The feedback winding  161  (with of turns) picks up a reflected voltage whose amplitude is in proportion to the secondary winding voltages of the transformer. After processed by the signal conditioning block  162 , the resulting feedback signal coupled to the error amplifier (E/A) is proportional to the output voltages. In one preferred embodiment, the functions performed by the block  162  include rectification and filtering, and possibly some signal conditioning to remove the effect of noise and irregularity caused by switching actions, By making the error/amplifier&#39;s reference voltage proportional to the input voltage with the right scaling factor M, all resonant tanks should have a theoretic voltage gain of close to one, so the effective voltage gain of the equivalent resonant tank representing the multi-tank system in this converter is approximately 1 in steady state. Considering the parameter tolerance in the resonant tanks, the reference, and the feedback control, the steady-state switching frequency may not be exactly the intended resonant frequency of the resonant tanks, or the same as the actual resonant frequency of any resonant tank. However, it will not deviate significantly from the intended resonant frequency if the component tolerance is not too high, for example not higher than +/−20%. This mode of operation is called controlled near resonance operation. As a result, the converter will automatically operate at a frequency very close to (for example within 10%) the resonant frequencies of all resonant tanks, achieving good efficiency and good output voltage regulation without sensing each output. Please note that in other embodiments this control scheme can work well also by sensing the output voltage directly at one of the outputs (instead of through a feedback winding), but such control system with direct sensing is more complex and expensive due to the need of signal isolation. 
     The existence of a closed-control loop in the controlled near resonance operation also allows the transient performance be controlled if necessary. Especially, for large transients such as startup or fault protection, a soft start or soft transition can be controlled through duty cycle or switching frequency of the converter, or through trajectory control of the resonant tank variables (the current in a resonant inductor, transformer winding or switch, the voltage across a resonant capacitor, or both). 
       FIG. 14( b )  shows the simulated waveforms of the converter in  FIG. 14( a )  in steady state operation. The primary switches Sp 1  and Sp 2  are gated on and off in complementary mode, with each having about 50% duty cycle. A short transition period is added between a switch&#39;s turn-off and the other switch&#39;s turn-on, to avoid cross conduction and allow zero-voltage turn on. It can be seen that zero voltage turn-on has been achieved for both Sp 1  and Sp 2 . However, in other embodiments the resonant converter can be operated to achieve zero-current turn-off of Sp 1  and Sp 2 , as is well known in the industry. Usually, to achieve zero voltage turn-on the magnetizing inductance of the transformer T is intentionally reduced. It can been seen also that the amplitude of the reflected voltage from the feedback winding nf is very close to an output voltage with same number of turns as nf. The small difference is mainly due to the power losses in the output circuit. Such deviation can be reduced if desired by considering power loss related information such as the transformer or switch current in the feedback or reference circuit of the error amplifier denoted as E/A in the regulation circuit  165 . Although synchronous rectifiers S 1  through S 4  are shown in the secondary in  FIG. 14( a ) , diode rectifiers can also be used to replace any or all of the synchronous rectifiers, as usual in power converters. The gate drive signals of synchronous rectifiers can be obtained from control signal in the primary through a gate drive transformer or other isolation components, or from windings on the main power transformer T, or from the current signals in the associated secondary windings or from the currents in the devices themselves. More outputs can be obtained by adding more secondary windings and associated resonant tanks and rectifier circuits. 
     This topology can also be used to operate away from the resonant frequency and thus achieves different voltage gains to regulating the output voltages to desired values. As long as the equivalent LC values in each resonant tank (considering the scaling factor of the transformer&#39;s turns ratios) are approximately the same, the double side resonance can achieve good cross regulation between the outputs. 
     The main power switches (usually power MOSFETs) should be designed to have low gate charge, so the driving loss is not significant. This means that the gate threshold voltage of the switch and the drive voltage applied on the gate should be reduced whenever possible. If the switching frequency used is high, it may be desirable to use resonant gate drive technologies to further improve the power converter efficiency. For a converter running around resonant frequency, it&#39;s also possible to use similar timing and drive circuit of the primary power switches (in this example Sp 1  and Sp 2 ) to drive the synchronous rectifiers, making the converter design much easier.  FIG. 15( a )  shows an example of a resonant gate drive circuit. A bridge circuit consisting of drive switch Network  180  (S 1  through S 4 ) converts the control signal for a controller to an ac waveform. The ac waveform is fed into a resonant circuit coupled to a gate drive transformer T 1 . Cp in the primary winding of T 1  can serve for dc-blocking, or be part of the resonant capacitance, or be used for both purposes. Lp, Ls 1 , Ls 2 , and Lsx are the resonant inductors, and each of them can be a discrete inductor, the leakage inductance of the corresponding transformer winding, or any combination thereof. Cg 1 , Cg 2  and Cgx are the equivalent gate-source capacitance of the switches to be driven, or any additional capacitance put there, or any combination thereof. Cg 1 , Cg 2 , and Cgx are also part of the resonant capacitance. To make sure that each drive output has similar performance, it&#39;s important to make sure that each component associated with each drive winding of transformer (n 1 , n 2 , nx) has similar equivalent value to its counterpart in any other driver winding when transferred to the primary side. If different numbers of turns are used in the driving windings, then in any secondary winding circuit, the secondary inductance (Ls 1 , Ls 2 , and Lsx) should be approximately proportional to the square of the number of turns of the winding, while the capacitance should be approximately inversely proportional to the square of the number of turns of the winding. If the equivalent gate-source capacitance of any switch is different from another&#39;s, external capacitors of appropriate values can be used to make the equivalent capacitance reflected to the primary approximately the same for all drive windings. In this way, all the resonant components can be grouped into a series resonant tank of inductance Lr (including the leakage inductance of the transformer and any trace and packaging inductance) and resonant capacitance Cr. Traditionally, a resonant gate drive involves one resonant state in one direction (charging and discharging), so the gate voltage is basically sinusoidal. Issues with this kind of driving scheme is that it takes a relatively long time for a switch to change from OFF state to ON state or vice versa, and the gate drive waveform varies strongly with the variation of the inductance and capacitance values. As a result, the converter performance is compromised. To improve upon this, a new composite driving scheme can be used.  FIG. 15( b )  shows the concept of the composite driving technique in state plane. As is well known, a resonant tank&#39;s normalized inductor current i(Lr) and the normalized capacitance voltage moves in a circular fashion in a resonant state. With the composite driving technology, the resonant tank operates in more than one state in the charging phase and discharging phase of a switching cycle. 
     In  FIG. 15( b )  two states are shown for each direction. The first state is shown by the big arcs in the trajectory, during which the voltage applied to the resonant tank is zero, and the resonant voltage (and thus gate voltages) changes polarity, as the center of the arcs is (0, 0). This can be called the polarity reversing state. The other state is shown by the small circles, during which the drive power supply Vcc is applied to the resonant tank in positive or negative direction, so the centers of the circles are (+Vcc, 0) and (−Vcc, 0) respectively. Because the radius of the small circles is much smaller than Vcc, during second state the resonant voltage fluctuate around Vcc or −Vcc slightly, so roughly the gate voltage can be considered to be pseudo “clamped” around Vcc or −Vcc with a small ripple. This state can be called a pseudo clamped state. Because a trajectory can circle the center in second state (the pseudo clamp state) as many times as required, the gate drive voltage can stay in the pseudo “clamped” mode for a relatively long and controllable time which is terminated by the start of the polarity reversing state in the other direction. 
     A complete switching cycle can be divided into 4 phases. The following is an example for explanation: 
     Assuming at the start the drive circuit is in positive charging phase with S 2  and S 4  turned on (S 1  and S 3  are off), so the circuit is in positive pseudo clamp state. The drive outputs Drive  1  and Drive X have positive voltages around a maximum value determined by Vcc with a scaling factor associated with the winding turns ratios and the ratios of resonant capacitances. The switches coupled to Drive  1  and Drive X are turned on. The other drive output Drive  2  has a voltage around negative maximum, so the switch coupled to it is off. The drive circuit and the switches will be in this state until S 2  is turned off and S 1  is turned on. 
     After S 2  is turned off and S 1  is turned on, the drive circuit enters into negative polarity reversing state, and the resonant voltage and thus the gate voltages cross zero and reverse their polarities. 
     When the resonant capacitor voltage reaches close to −Vcc (or the current in Lr is about zero), S 4  is turned off and S 3  is turned on, now the drive circuit enters the negative pseudo clamp state, and the gate voltages are in pseudo clamped mode again, but the voltage polarities are versed compared to the starting point mentioned above. This state terminates until S 1  is turned off and S 2  is turned on. 
     After S 1  is turned off and S 2  is turned on, the resonant voltage and gate drive voltages start their journey to change polarity again in positive polarity reversing state. 
     When the resonant voltage V(Cr) reaches close to Vcc, S 3  is turned off and S 4  is turned on, the drive circuit enters into the positive pseudo “clamp” state, and the system finishes a full switching cycle. 
     By adjusting the relative duration of the pseudo clamp mode, the drive circuit&#39;s operation is more independent of the resonant tank parameters, and the transition time between a switch&#39;s states can be faster than in traditional control. Moreover, the duty cycle of the main switches can even be adjusted as the durations of pseudo clamp modes do not have to be equal in positive direction and negative direction. This gives better performance and freedom in the converter design. The key to this scheme is that the control timing of the drive switches S 1  through S 4  should be determined by comparing the resonant capacitor voltage to Vcc and −Vcc, or comparing the resonant inductor current to zero (or a small value), so this is basically a drive scheme with feedback. Please note that actual feedback signal, either the resonant capacitor voltage or the inductor current, doesn&#39;t have to be measured at the primary side of the transformer T 1 . As long as the proper scaling is considered (due to the transformer turns ratios and the LC component values), the feedback signal can be taken from one or more driving circuit on the secondary side of the transformer, or from one or two feedback windings on the transformer. Alternatively, the duration of the polarity reversing states can be fixed according to the resonant tank LC parameters (roughly half the resonant cycle). This is a simple open loop control, but should work reasonably well for many applications. Also, if more drives are needed, more driving windings and associated circuit can be added as this drive structure is easily expandable. 
     The above schemes are relatively easy to implement with modern analog and/or digital electronics. The above explanation is for steady state operation. During transients, such as during a startup, the control should be slightly modified. For example, during a startup, the switch frequency and/or duty cycle should be changed to reduce voltage and current stresses in both the drive circuit and the main power circuit. One possibility is to keep the duration of the polarity reversing state about the same as in steady state (without comparing the capacitor voltage), but gradually increase the duration of pseudo clamp states from zero to normal value in a soft-start process. Other control methods such as switching frequency control, duty cycle control, and trajectory control can also be used to limit the stress level. For fault protection, hick-up schemes with shutdown and soft-start phases can be considered. 
     The above circuit many have a draw back for some applications: a gate drive voltage swings to both positive and negative value symmetrically, and its dc component is zero. It may be desirable to move the gate drive voltage to more positive or more negative in certain applications, for purposes such as adjusting the dead-time between switches.  FIG. 15 ( c )  shows an implementation with voltage shifting and voltage clamping techniques. Diodes D 1  and D 2  create a positive dc voltage for Drive  1  and Drive  2 , so the voltage waveforms will be shifted up for both drive outputs. If a negative bias is desired in any output, the polarity of the associate diode can be reversed. The optional resistors R 1  and R 2  are used to discharge the gate voltages. By adjusting the values of Cs 1  and Cg 1 , the waveform of Drive  1  voltage can be shifted up and down. Similarly, the voltage of Drive  2  can be shifted by the values of Cs 2  and Cg 2 . The switch Sg 3  clamps the negative voltage of Drive  3  to about zero. When the clamp switch is turned on, the resonant capacitor it clamps is shorted, and totally resonant capacitance of the resonant tank is changed. As a result, the voltage waveform may be changed. The capacitances of Cs 3  and Cg 3  and the number of turns of winding n 3  can be optimized so the effect of this capacitance change is reduced. As an alternative, if needed another clamp switch can be put in parallel with Cs 3 , with its gate connected at the drain of Sg 3 . With the symmetric connection of these two switches, if Cs 3  and Cg 3  have the same capacitance, then the equivalent capacitance in the resonant tank doesn&#39;t see much change during the different phases of operation, and thus the gate voltage waveform is more consistent. Such a symmetric arrangement of clamp switches is shown with the circuit coupled to winding nx. The driving winding nx in this figure powers two drive outputs coupled together. These two outputs have reverse polarities, and can be used to drive for example two bottom switches in a bridge configuration or the power switches in a push-pull configuration. The optional clamp switches Sgx 1  and Sgx 2  can clamp the negative gate voltages to zero if needed. As the clamp switches is turned on alternatively, the resonance of this resonant tank is not affected much by the voltage clamp. More drive outputs can be added by repeating the circuit patterns in  FIG. 15( c ) , and the voltage shifting and voltage clamping techniques shown in the figure can be used in any combination as needed. 
       FIG. 15( d )  shows key simulated waveforms of the circuit shown in  FIG. 15( c ) , and verifies the operation principles discussed above. 
     The above discussed gate drive technologies can be used to drive power switches and synchronous rectifiers in different topologies. The controller can take the feedback information of resonant capacitor voltage or transformer current to accomplish closed-loop driving, or use a configurable delay corresponding to the polarity reverse state to accomplish open loop driving. 
     The topology in  FIG. 14( a )  uses one driving winding for each output voltage. To simplify the implementation, the output voltages may be combined into voltage pairs as needed by a power synthesizer.  FIG. 16  shows a practical implementation. Now the resonant capacitors in the secondary side of the transformer Cr 1 , Cr 2 , cr 3  and Cr 4  are arranged to be in series with the secondary windings, and will resonate with the leakage inductance of their respective secondary windings. Since the leakage inductance of a secondary winding is proportional to the square of the number of turns of the winding, the resonant capacitance coupled to each winding should have a reciprocal relationship to the square of the number of turns of the winding so that all the resonant tanks can resonate at the same frequency, i.e.: 
     
       
         
           
             
               
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     If needed, discrete resonant inductors can be put in series with the transformer windings, and the value of resonant capacitors should be adjusted accordingly. 
     To simplify the transformer structure, the secondary windings in each voltage pair circuit is connected together: n 2  is connected with n 1 , while n 4  is connected with n 3 . Please note that the polarity of n 2  is reverse to n 1 &#39;s polarity, and the polarity of n 4  is reverse to n 3 &#39;s. This reverse arrangement of windings allows the rectifiers (synchronous rectifiers or diodes) to be connected together in a dual totem network  181  and  182 , as is shown in  FIG. 16 . The outputs are arranged in asymmetric pairs  191  and  192 . With this arrangement, when Sp 1  conducts, synchronous rectifiers S 11 , S 22 , S 31  and S 42  should be gated on. When Sp 2  conducts, synchronous rectifiers S 12 , S 21 , S 32 , and S 41  should be gated on. Because now the top winding in a voltage pair circuit charges both output voltages in the pair, to maintain a doubling relationship between the output voltages in the pair, the top winding should have three times the number of turns as the bottom winding: n 2 =3*n 1 , n 3 =4*n 1 , n 4 =3*n 3 . 
     The doubling relationship of output voltages can also be realized with simpler winding structure by duty cycle control. In the half-bridge topology shown in  FIG. 17( a ) , multiple windings are coupled to the same magnetic component (transformer T 1  in this case), where there exists a relationship between the numbers of turns in the windings: n 3 =4*n 2 . Sp 1  and Sp 2  are controlled in complementary mode but with different duty cycles. The duty cycle of Sp 1 , D 1 , is twice the duty cycle of Sp 2 , D 2 : D 1 =2*D 2 . If the short transition periods in which both switches are off are ignored, then we have: 
               D   ⁢           ⁢   1     ≈       2   3     ⁢           ⁢   and   ⁢           ⁢   D   ⁢           ⁢   2     ≈       1   3     .           
Then by virtue of volts-second balance of the magnetic component T 1 , the output voltages in steady state have a doubling relationship:
 
     
       
         
           
             
               
                 V 
                 ⁢ 
                 
                     
                 
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                 * 
                 
                   
                     n 
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                     2 
                   
                   
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     In this way, one secondary winding is used to produce a pair of outputs. In one embodiment, the two output voltages are different in a pair. In other embodiments the two voltages can be made to have a different relationship by adjusting D 1  and D 2  differently. The synchronous rectifiers S 1  through S 4  can be replaced by diodes if needed. The gate drive signals for the synchronous rectifiers can be in synch with the primary switches, or emulate ideal diodes: gated on when the body diode starts conducting, and gated off when the current through it is lower than a threshold value. The drive signals can be obtained from primary control signal or through transformer windings of T 1 , or through current signals in the transformer windings or the switches. As an option, the above discussed composite resonant driving technology can also be used, as the durations of the pseudo clamp state can be set to different values during Sp 1  conducting phase and Sp 2  conducting phases. 
     In practice, closed-loop control may be required to maintain the voltage relationship. A possibility is to use voltages in the primary side, such as capacitor voltages or a winding voltage to simplify the control cost. For example, when the voltage of Cp 1  is less than 50% of the voltage across Cp 2  (or less than ⅓ of the dc-link voltage across the half-bridge configuration of Sp 1  and Sp 2 ), then the duty cycle of Sp 1  (D 1 ) should be adjusted lower. When the voltage of Cp 1  is higher than 50% of the voltage across Cp 2 , then the duty cycle of Sp 1  (D 1 ) should be adjusted higher. A feedback controller can automatically adjust the duty cycle to the right value. 
     Due to current-fed nature of this topology caused by the input inductor L 1 , a snubber circuit or clamp circuit may be used to reduce the voltage stress of primary switches. In  FIG. 17( a )  an optional passive snubber consisting of Cd and Rd is inserted across the dc link and between the input inductor L 1  and the switches. Other types of snubber or clamp circuits can also be used.  FIG. 17( b )  shows simulated waveforms of this topology. It can be seen that Sp 1  is turned on with reduced voltage, and Sp 2  is turned on with zero voltage. The ability to achieve soft-switching in this topology improves its efficiency. 
     The topology in  FIG. 17( a )  has limited ability to adjust the output values by adjusting the duty cycles of the switches. If a wide range of adjustment is needed, a regulation stage can be added at the input.  FIG. 18  shows a buck regulation stage  195  consisting of buck switch Sb 1  and synchronous rectifier Sd 1  is added. The switching of Sb 1  can be synchronized with Sp 1  and Sp 2 , or can be arranged independently. 
     Different implementations can be obtained by changing the half-bridge configuration in  FIG. 17( a )  to different topologies. A full-bridge configuration using the same duty cycle control principle is shown in  FIG. 19  which can two pairs of outputs. The secondary windings also have a quadruple relationship between the number of turns: n 3 =4*n 2 . The primary switch network consisting of four primary switches Sp 1  through Sp 4  are operated in 3 states alternatively: 
     State 1: Sp 1  and Sp 3  are turned on, and Sp 2  and Sp 4  are turned off. Secondary side synchronous rectifiers S 1  and S 3  conducts, and S 2  and S 4  are gated off. The transformer winding voltages are determined by V 1  and V 3 ; 
     State 2: Sp 1  and Sp 3  are turned off, and Sp 2  and Sp 4  are turned on. Secondary side synchronous rectifiers S 1  and S 3  are gated off, and S 2  and S 4  are gated on. The transformer winding voltages are determined by V 2  and V 4 ; 
     State 3: all primary switches Sp 1  through Sp 4  are turned on (or at least switches in one of the two legs, i.e. Sp 1  and Sp 2  or Sp 3  and Sp 4 , are turned on), all secondary synchronous rectifiers S 1  through S 4  are gated off. The transformer winding voltages are zero. 
     In a switching cycle these states may have different sequences, but a preferred sequence is that an active state (State 1 or State 2) is followed by a zero state (State 3), and then the other active state (State 2 or State 1), and then a zero state (State 3). The duty cycle of State 1 (D 1 ), should be kept at twice the duty cycle of State 2 (D 2 ): D 1 =2*D 2 . The remaining duty cycle D 3 =1−D 1 −D 2  is the duty cycle of zero state S 3 , and should be split into two parts to be inserted between D 1  and D 2 . In steady state, the ideal output voltages are: 
     
       
         
           
             
               
                 V 
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                 V 
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     Therefore, by controlling D 1 , the output voltages can be adjusted to the desired value. This control can be implemented as a current mode or voltage mode closed-loop control if precise regulation is required. 
     To maintain the doubling relationship between V 2  and V 1 , another closed-loop control may be used to adjust the relative value of V 2 . If V 2  is higher than the ideal value (2*V 1 ), D 2  should be increased. If V 2  is lower than the ideal value, D 2  should be reduced. 
     If the duty cycle of State 3 (zero state) is intentionally kept close to zero, then this converter becomes a multi-output bus converter, which may have high efficiency. 
     Because the primary side switches are not clamped, there may be high voltage stress during a switch&#39;s turn-off. Proper measures, such as a snubber or an active clamp circuit can be used to reduce the voltage stress. In  FIG. 19 , a snubber/clamp circuit  194  is shown. In a preferred embodiment, the circuit  194  may be a passive snubber consisting of a capacitor and a resistor. In another embodiment, the circuit  194  may have one or more active clamps. 
       FIG. 20  shows the simulated key waveforms of the converter shown in  FIG. 19  in an example implementation, and verifies the operation principle discussed above. Although doubling relationship is shown in this embodiment for the voltages in a pair, other relationships are possible by changing the relationship between D 1  and D 2 . If D 1 =D 2 , then the two voltages in pair  191  or  192  will be equal. 
     In some power applications such as smart phones and other mobile devices, there is no isolation needed between the input and the outputs of the power supply. Then a buck converter based multi-output configuration, with similar principles as discussed before, can be used to generate the input voltages for the power synthesizer.  FIG. 21  shows two more examples. In  FIG. 21( a ) , flyback windings n 2 , n 3  and n 4  are added to the magnetic component (inductor T 1 ), and S 2  through S 4  work as synchronous rectifiers (in rectifier networks  181  and  182 ). When S 1  conducts (S 1   d  is gated off) in the buck switch network  180 , switches in the rectifier networks  181  and  182  (S 2  through S 4 ) are gated off. When Sd 1  conducts (S 1  is turned off) in the freewheeling mode of the buck converter, S 2  through S 4  are gated on so power is transferred from V 1  to V 2  through V 4 , and the relationship between the output voltages is determined by the turns-ratios of the windings. The circuit in  FIG. 21( b )  works similarly to the circuit in  FIG. 21( a ) , but the rectifier networks  182  and  181  are arranged slightly differently. 
     The power synthesizer described above has a minimum voltage of zero. If the power amplifier needs a higher minimum voltage, another voltage source can be added to the power synthesizer.  FIG. 22  shows an example of a fast dynamic power supply based on a buck-based topology. V 1  is the minimum voltage applied to the PA and is generated by the buck switch cell  180  directly. Through using windings n 2 , n 3  and n 4  with proper number of turns, V 2 , V 3  and V 4  can have the correct voltages, and are fed into an eight-level power synthesizer consisting of a two-level cell  140  and 1 four-level cell  150 . With this arrangement, eight different voltage levels can be achieved. 
     Please note that in all the above descriptions, the input power source Vin can be a primary power supply such as a battery pack, a battery plant, a photovoltaic cell, a fuel cell, or the output of a power converter such as a dc-dc converter, or an ac-dc converter. When a dc-dc converter or an ac-dc converter is used, the input voltage Vin can be dynamically adjusted. The input voltages to the power synthesizer (V 1 , V 2 , and so on) can be adjusted dynamically through changing Vin or controlling the switches in the multi-output power supply to optimize the performance (such as the efficiency) of the whole system. For example, when the system transfer power is increased due to heavier usage of users in a base station, the voltages can be increased. When the needed transfer power is reduced, the voltages can be decreased accordingly. For ET systems, the output voltages of the power supply can be controlled with a relatively slow voltage control loop (compared to the frequency of power synthesizer, or the envelope spectrum of the signal processed by the power amplifier) and follow the envelope peaks which are related to the peak or average power of the power amplifier&#39;s output over a duration of time, and change relatively slowly. Such a system is sometimes called average mode ET control. 
     Please note also that although in this disclosure multi outputs are used to describe the power supply topologies and control schemes, the concepts and principles can also be used for single output converters with straightforward modification (such as remove unused secondary windings and their associated circuitry). Of course, single-output power supply may not need the double-side resonance technique if the output voltage is sensed directly at the output. However, sensing the output voltage through a feedback winding at the primary side can still work well with double-side resonance technique even for the single-output applications, and using the feedback control to automatically operate a resonant converter at or near its resonant frequency is still a very effective technique for single-output converters. 
     One aspect to improve the performance of a power converter is to limit the range of the switching frequency. This is very important especially in resonant power converters and resonant power supplies.  FIG. 23  shows a technique to accomplish this purpose. In an embodiment of a resonant power converter, an input voltage circuit  155  generates an input voltage Vin. The circuit  155  may be an acdc power converter connected to an ac power source, or a power terminal or power bus to bring in an external dc source. Vin may vary in a wide range due to various reasons, such as the voltage ripple in the output voltage of an acdc converter or diode rectifier. A switch network  180  converts Vin into an ac or pulse waveform, which is fed into a primary resonant tank  183 . Optional transformer and secondary resonant tank  185  are coupled to the primary resonant tank  183  and output circuit  182 , which generates one or more output voltages or currents. A frequency control block  161  controls the frequency of the switch network  180 , and thus controls the operation of the resonant power converter  2300  and may regulate its output voltage, current or power. Resonant tanks  183  and/or  185  may have more than one resonant modes. For example, one of a resonant capacitor or inductor in a resonant tank may be variable and take different values, so that the resonant frequency and/or characteristic impedance of the resonant tank can change in a certain range, creating different resonant modes with different operating characteristics in the operation of the resonant power converter. A resonant mode selection circuit  162  is used to select a resonant mode according to the operating conditions and requirements. For example, one objective of the resonant mode selection block  162  may be to limit the switching frequency range of the resonant power converter so its design can be optimized. To do this, block  162  may use the input voltage as an input to choose a suitable resonant mode for a given range of input voltage. If the input voltage moves out of this range then a different resonant mode is chosen which is more suitable for the changed input voltage. Alternatively, block  162  may use information from the frequency control block  161  as an input. When the frequency control requires the switching frequency of the resonant power converter to enter a undesired range (for example, to go higher than a upper limit or lower than a lower limit), the resonant mode selection block  162  changes the resonant mode so the switching frequency will remain in the desired range. In this way, even if Vin may change in a wide range due to a line-frequency ripple or other reasons, or an operating characteristics may change in a wide range which may cause the switching frequency to change in a wide range, by switching among different resonant modes properly the resonant mode selection block  162  can maintain the operating of the resonant converter within a narrower switching frequency range, which may result in lower cost design and/or better performance of the resonant converter. Of course, the resonant mode selection block  162  may have other objectives, such as to increase the efficiency of the resonant power converter. This switched resonant mode operation can be implemented in primary resonant tank  183 , secondary resonant tank  185 , or both. 
       FIG. 24  shows an example to implement the technique of switched resonant mode described above. Sp 1  and sp 2  form a half-bridge switch network  180 . Secondary circuit  186  is not shown in detail for the sake of brevity, and may consist of any suitable circuit to produce one or more suitable outputs. Lrp, which may include the leakage inductance of transformer T, is a resonant inductor. Cx 0 , Cp 1  and Cp 2 , as well as the switchable capacitor network  163  consisting of Cx 1 , Cx 2 , Sx 1  and Sx 2 , form equivalently a resonant capacitor. By controlling the ON/OFF states of Sx 1  and Sx 2 , four different capacitances, including 0, Cx 1 , Cx 2 , and Cx 1 +Cx 2 , can be put into parallel with Cx 0 , resulting in 4 different resonant capacitances for the resonant tank  183 . This is equivalent to change the capacitance of the equivalent resonant capacitor. In this way, 4 resonant modes are generated, and can be switched by resonant mode selection block  162 . 
     In low power application, a single-end topology such as a flyback, forward, or class-E resonant converter may be more desired due to lower implementation cost.  FIG. 25  shows a single-end resonant topology which is a hybrid of a flyback converter and a class-E converter. Sp 1  forms a primary switch network  180 , Cp 1  and the primary winding of transformer T ( 184 ) form a primary switch network  183 , Cs 1  and a secondary winding of transformer T form a secondary resonant tank  185 . Cp 1  may include the output capacitance of power switch Sp 1 . The transformer T may be implemented on a PCB (printed circuit board) or multiple PCBs to have relatively consistent magnetizing inductance and leakage inductance. Multiple PCBs may be separated to increase safety distance or for other system consideration. Of course, separate inductors may be added to the primary winding or secondary winding of the transformer, and different resonant tank configurations, such as a parallel resonant or parallel-series resonant tank, may be used for the primary or secondary resonant tank. A half-bridge switch network of S 11  and S 12  and an output capacitor Co form an output circuit  182  to produce an output Vo. If desired, a full-bridge switch network may be used in the output circuit. Furthermore, multiple secondary resonant circuits and output circuits may be used to produce multiple outputs if needed. Through proper duty cycle and/or switching frequency control, Sp 1  can work in zero-voltage switching, and the converter can output an output voltage in a suitable range. 
     The switched resonant mode operation can be applied to this resonant converter by making Cp 1 , Cs 1 , or both changeable.  FIG. 26  shows an implementation example to change the equivalent capacitance of Cp 1 . In the embodiment of  FIG. 26 , a switchable capacitor network  163  consisting of Cx 1 , Cx 2 , Sx 1  and Sx 2  is put in parallel with Cp 1 . A resonant mode selection block  162  can use Vin or frequency information in frequency control block  161  to decide the status of Sx 1  and Sx 2 , and thus choosing the right resonant mode for the primary resonant tank, so a good performance can be obtained with a low implementation cost. Please note that the capacitance of Cs 1  in the secondary resonant tank  185  may also be changed dynamically. To reduce cost, it&#39;s better to adjust Cs 1  just according to a variable in the secondary circuit, such as an output voltage change or load change. 
     With the switching frequency range of a resonant converter reduced by the switched resonant mode technique, resonant gate drivers such as the ones described in  FIGS. 15( a ), 15( b ), 15( c ) and 15( d )  can be more easily used. For a single-end topology such as the one shown in  FIG. 26 , a simpler resonant gate drive topology shown in  FIG. 27  may be used. In this embodiment, switch network  180  consisting of switches S 1  and S 2  converts a dc drive voltage Vcc into a pulse waveform, and couples the waveforms to a resonant tank  183 , which consists of a resonant inductor Lg, and a resonant capacitor Cgs. Lg may consist of a parasitic inductance in the gate path, such as PCB trace inductance and/or bond wire inductance coupled between the switch network  180  and the gate and source of the main power switch, as well as any added discrete inductor. Cgs may include parasitic capacitance such as any PCB trace or Pad capacitance and the input capacitance of the main power switch (not shown in the figure) which is driven by this gate drive circuit, and may include any added discrete capacitor. Switch S 3  is optional and may clamp the gate-source voltage and ensure the turn-off of the main power switch, as will be discussed later.  FIG. 28  shows a phase diagram of an intended operation of the gate drive circuit in  FIG. 27 . Similar to the drive circuit shown in  FIG. 15( a )  and the phase diagram in  FIG. 15( b ) , the resonant gate drive circuit shown in  FIG. 27  may utilize the concept of composite driving technique where the resonant tank in the gate drive circuit may operate in more than one states in the charging phase and discharging phase of a switching cycle. 
     The gate drive circuit&#39;s operation can be explained more clearly with the help of the phase diagram shown in  FIG. 28 . We can investigate first the operation of the circuit without S 3 . Assuming initially the circuit is at the following condition:
 
 Vgs= 0, Ig≈Vcc/Zo , where  Zo  is the characteristic impedance of the resonant tank, i.e. the square root of ( Lg/Cgs ).
 
     This corresponds to point A in  FIG. 28 . To initiate a turn-on process for the main switch, S 2  should be turned off, and S 1  should be turned on. Then the state variable (Vgs, Ig) of the resonant tank  183  will move clockwise along the arc GAB in the phase diagram. The arc GAB is part of a circuit (the resonant circle) with its center at (0, 0) and a radius of about Vcc. This is a resonant state, and during this state, Vgs gradually increases, while Ig gradually decreases, both in a sinusoidal fashion. When Vgs exceeds the gate threshold voltage of the main power switch, the main power switch will turn on. When Vgs approaches Vcc, for example Vgs reaches a level of 90% of Vcc, S 1  is turned off and S 2  is turned on. This corresponds to point B in  FIG. 28 . The duration of the resonant state is a little lower than ¼ of the resonant period determined by the values of Lg and Cgs. After this, the state variable (Vgs, Ig) of resonant tank  183  will enter a small arc BCD or a small circle BCDEB in the phase diagram, with the center of the both at (+Vcc, 0). Because the radius of the small circle or small arc is much smaller than Vcc, during this state Vgs fluctuates around Vcc slightly during this mode (for example staying within +/−30% of the average value during this mode), and Ig fluctuates around zero slightly, so roughly the gate voltage of the main power switch can be considered to be pseudo “clamped” around Vcc with a small ripple, which can be designed to be less than a limit suitable for driving the power switch, say within 30% of the average voltage during this state. This state is a pseudo clamped state. Each circle corresponds to a resonant period in time. Because in this state the state variable can circle the center (Vcc, 0) as many times as required, the gate drive voltage can stay at a level suitable for driving a switch in the pseudo “clamped” mode for a relatively long and controllable time, which is terminated by the start of the turn-off phase at point D. Therefore, by adjusting the duration of the pseudo clamped mode, the on time of the main switch can be controlled, and thus the duty cycle and/or the switching frequency of the power converter can be controlled. Point D is a mirror to Point B along Vgs axis, at which Vgs is the same as at Point B, but Ig changes direction compared to at point B. At this point S 2  is turned off and S 1  is turned on, so the state variable of the resonant tank  183  will moves clockwise along the arc DFGA in the phase diagram shown in  FIG. 28 . Ideally, the arc DFGA is part of the resonant circle with its center at (0, 0) and a radius of about Vcc. During the first part of this resonant state, Vgs gradually decreases, while Ig gradually increases, both in a sinusoidal fashion. When Vgs moves below the gate threshold voltage of the main power switch, the main power switch will turn off After this, the state variable of the resonant tank  183  continues to move along the arc DFGA, with Vgs and Ig both changes polarity along the course, and reaches point A at the end. The time duration of the resonant state from point D to point A is about ¾ of a resonant period. Then next switching cycle begins repeating the above process. 
     With the circuit in  FIG. 27  and the phase diagram in  FIG. 28 , the duration of main power switch&#39;s OFF state is relatively fixed. By adjusting the relative duration of the pseudo clamp mode, the drive circuit&#39;s operation is more independent of the resonant tank parameters, and the transition time between a switch&#39;s states can be faster than in traditional control. Moreover, the duty cycle of the main switch can even be adjusted in certain degree. A potential shortcoming is that the main switch&#39;s turn-off time and duration may not be fully controlled. To improve this, S 3  can be add as shown in the dotted line in  FIG. 27 . The operation of the circuit with S 3  connected is basically the same during the turn-on process and the first part of the turn-off transition of the main power switch, during which S 3  is turned off. The difference starts when the state variable of the resonant tank  183  reaches point F. At point F, due to conduction of the body diode or anti-parallel diode of S 3 , the state trajectory will stop movement, so Vgs is clamped at zero, with Ig equal to −Vcc/Zo. This is a first gate freewheeling state, during which S 3  may be gated on to reduce the power loss. Ignoring the conduction loss of the drive current path, this state can be as long as forever. In a practical operation, even with the conduction loss considered this state can last a long enough time to allow a good duty cycle control of most power converters. When it is desired to start a turn-on process for the main power switch, S 1  can be used off and S 2  can be turned on possible. Please note that S 2  may be turned on with zero-voltage as the negative Ig can force the body diode of S 2  to conduct before S 2  is gated on. During this state of the operation, the state variable of the resonant tank  183  moves along the dotted straight line FOA in the phase diagram. During this state, Vgs is clamed to zero by S 3 , but Ig is charged linearly by Vcc, so this stage can be called inductor charging stage. If S 3  was not gated on previously, it can be gated on before Ig reaches zero. Also, during this stage if S 2  is turned off and S 1  is turned on, the gate drive circuit can enter into second gate freewheeling stage in which Vgs is clamped zero, and Ig doesn&#39;t change. If Ig is close to zero, then the power loss in the gate drive circuit is very small in this state. Therefore, it may be desirable to use this second gate freewheeling state alone or in combination with the first gate freewheeling state to control the turn-off time of the main switch. When the state variable reaches point A, S 2  and S 3  can be turned off, and S 1  can be turned on. The positive Ig makes it possible to turn on S 1  with zero-voltage. After this, the resonant gate drive circuit enters the resonant stage in the turn-on process as discussed above. 
     At any time the main switch needs to be turned off such as in a faulty condition or shut-down operation, S 1  and S 2  should be turned off and S 3  can be turned on. In this way, the turn-on of S 3  can ensure the turn-off of the main power switch. Similar turn-off mechanism can be applied to the gate drive circuit shown in  FIG. 15( c ) , where with proper circuit parameters, Sg 3  in secondary circuit block  186  can reduce the output gate voltage of Drive  3  to close to zero when the switch network  180  produces zero voltage, i.e. when both S 1  and S 4  (or alternatively both S 2  and S 3 ) are turned on. If similar circuit is used for other drives, all main power switches may have a long turn-off state, which allow a power converter to be in a shut-down or fault-protection mode. 
     Another consideration is the start-up of a resonant gate drive circuit. It is interesting to know that the state trajectory during the resonant states in both the phase diagrams shown in  FIG. 15( b )  and  FIG. 28  is part of a resonant circle with its center at (0, 0) and a radius of about Vcc. Since the initial condition of the state variables of the resonant gate drive circuit is (0, 0), the gate-drive start-up can start with Vcc being applied to the resonant tank in the circuit. In the circuits shown in  FIGS. 15( a ) and 15( c ) , it means turn-on of S 2  and S 4  while turn-off of S 1  and S 3 . In the circuit shown in  FIG. 27 , it means turn-on of S 2  and turn-off of S 1  and S 3  (if it is connected). This will force the state variables of the gate drive resonant tank to move a long a start-up circle with its center at (Vcc, 0) and a radius of Vcc in the phase diagram. This start-up circle will cross the resonant circle after about ⅛ resonant period. At the cross point, Vgs is about 70% of Vcc. Therefore, the cross point can be determined by timing the duration of start-up state, or by comparing Vgs to a value close to 70% of Vcc. At the cross over point, the gate drive start-up process is over, and the status of switches in a gate drive circuit can be changed to the corresponding switch status in the corresponding resonant state as discussed above. According to this gate-drive start-up scheme, it only needs about ⅛ resonant period of the drive resonant tank to bring the gate drive circuits into normal operation. Once the gate drive circuit can start normal operation, the switch frequency and/or duty cycle of main power switches should be changed to reduce voltage and current stresses in a soft-start process, as discussed previously. 
     The key to the above drive schemes is to accomplish correct control timing for the drive switches S 1  through S 4 . These switches&#39; ON and OFF instants can be timed according to the duration of each state as discussed previously. However, when the resonant inductor and resonant capacitor, and thus the resonant period, have relatively big tolerance, it is difficult to do such fixed timing without careful measurement or calibration of the resonant elements or resonant period. Alternatively, the correct timing may be determined by comparing the resonant capacitor voltage to given reference points (such as a value close to +Vcc or −Vcc), and/or comparing the resonant inductor current to given reference points, such as zero or Vcc/Zo, so this is basically a drive scheme with feedback. The measurement of capacitor voltage is relatively straight-forward. The measurement of resonant inductor current (such as Ig) can be done on a resistor in the gate drive path, or through current mirrors of the gate drive switches. With modern analog and digital techniques, the signal sensing and processing according to the above schemes are all possible, and the sensitive part of a gate drive circuit may be implemented on an integrated circuit. 
     This resonant gate drive circuit is actually a resonant power converter. The circuit and technique discussed here can also be used in resonant power converters with straightforward modifications to suit the output requirements of the converter. 
     Because the dynamic power supply has high bandwidth requirements, the power synthesizer needs to work at very high frequency. Power integration techniques can be used to improve efficiency and reduce noise whenever possible. The switches are arranged in totem pole configurations in the synthesizer. In one embodiment, the two switches and their drive circuit in a totem pole are integrated together in a single die or a single package. In another embodiment, the four switches in a basic four-level cell and their drive circuits are integrated in a single die or package. In other embodiments, all switches of a power synthesizer and their gate drive circuit are better to be integrated into a single die, or into a single package. The control circuit of the synthesizer can be integrated too. It&#39;s preferable to include some bypass capacitors, such as C 2 , C 3  and C 4  in  FIG. 21  in the same package or the same die as the switches. If possible, the power supply switches (for example S 1  through S 4  and S 1   d  in  FIG. 21 ) can also be integrated into the same die or the same package. The drive circuit and control circuit of the multi-output power supply can be also integrated into the die or the package. In an embodiment, the active components and key passive components of the whole dynamic power supply can be integrated into a package, with one or more semiconductor dies and optional discrete passive components. Finally, the power amplifier itself can be integrated with the power solution in a single die, or in a multi-chip module package. In this way, the effect of package parasitics can be minimized, and optimum performance can be achieved. 
     The above discussion of the advanced power solutions is made mainly in the context of power amplifier applications, since power amplifiers are a very suitable target for high performance power technologies. However, the applications of the power technologies discussed are not limited to power amplifiers, and can be used in any other applications, devices and equipment which require high efficiency power conversion and power control. The power technologies can be used in stand-alone power devices, power chips, power modules and power supplies systems, or be used in devices or systems with power solutions as a built-in function. 
     Although embodiments of the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. 
     Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, composition of matter, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.