Patent Publication Number: US-2007123182-A1

Title: Local oscillator leakage cancellation in radio transmitter

Description:
FIELD  
      The present invention relates to local oscillator leakage cancellation in a radio transmitter.  
     BACKGROUND  
      In radio transmitters, a local oscillator (LO) is used to up-convert a modulated analog baseband or intermediate frequency signal to the final radio frequency (RF). All practical up-converters pass part, unintentionally, of the LO signal to their output. The LO may also leak in other ways to the transmitter output. The presence of an LO signal can in many ways be harmful to the transmitter, such as by generating switching transients in a TDMA transmitter or by extra loading of the power amplifier. In transmitter architectures based on an intermediate frequency, it is in principle possible to suppress the LO leakage by filtering. However, if the LO frequency is too close to the desired signal band, the filter requirements may become impractical. In direct conversion architectures, the LO is inside the transmitted signal bandwidth and needs to be cancelled in another way.  
      Since LO leakage depends on environmental factors, such as temperature and aging, in practice LO cancellation methods need to be adaptive. In the prior art, LO cancellation has been suggested to TDMA (Time Division Multiple Access) based transmitters. Then, LO parameters, such as leakage, may be estimated by comparing measurements on active and idle timeslots. The prior art methods are, however, not applicable to transmitters wherein transmission is continuous.  
     BRIEF DESCRIPTION  
      In one aspect of the invention, there is provided a radio transmitter, including means for up-converting an input signal by mixing the input signal with a local oscillation signal, means for extracting an observation signal from the up-converted signal, means for switching the observation signal between an ON state allowing throughput of the observation signal and an OFF state preventing throughput of the observation signal, means for down-converting the observation signal by mixing the observation signal with the local oscillation signal, means for filtering signal components around the down-converted oscillation signal, means for generating a compensation signal by using the filtered signal in the ON and OFF states of the observation signal throughput, and means for modifying the input signal with the compensation signal.  
      In another aspect of the invention there is provided a chipset, including means for up-converting an input signal by mixing the input signal with a local oscillation signal, means for extracting an observation signal from the up-converted signal, means for switching the observation signal between an ON state allowing throughput of the observation signal and an OFF state preventing throughput of the observation signal, means for down-converting the observation signal by mixing the observation signal with the local oscillation signal, means for filtering signal components around the down-converted oscillation signal, means for generating a compensation signal by using the filtered signal in the ON and OFF states of the observation signal throughput, and means for modifying the input signal with the compensation signal.  
      In still one aspect of the invention there is provided a method in a radio transmitter, including steps of up-converting an input signal by mixing the input signal with an oscillation signal, extracting an observation signal from the up-converted signal, switching the observation signal between an ON state allowing throughput of the observation signal and an OFF state preventing throughput of the observation signal, down-converting the radio frequency signal by mixing the radio frequency signal with the oscillation signal, filtering signal components around the down-converted oscillation signal, generating a compensation signal by using the filtered down-converted signal in the ON and OFF states of the observation signal throughput, and modifying the input signal with the compensation signal.  
      Preferred embodiments of the invention are disclosed in the dependent claims.  
      The invention relates to cancellation of LO leakage in a radio transmitter, such as a base station or a mobile phone. In the invention, the radio transmission is continuous, or there is at least a continuous leakage path between the transmitter LO input and the transmitter output. In the invention, the RF signal input to an observation receiver of the transmitter is switched periodically ON and OFF and the LO cancellation is based on the difference between the demodulator output in the OFF/ON states of the RF signal input. In hardware, this difference may be achieved by periodically inverting the output of the demodulator synchronously with the operation of the RF switch. Integrated quadrature demodulators may have differential outputs, and thus their polarity may be inverted by means of switches. In another embodiment, demodulator outputs may be sampled in an analog-to-digital converter (ADC) and polarity inversion may be performed in the digital domain.  
      The invention provides advantages, such as making effective LO cancellation possible in a transmitter using continuous transmission. Implemented digitally, the cancellation loop of the invention has the advantage of higher accuracy and lower cost, since the cancellation loop can then be integrated with the other digital circuitry in the TX (transmitter). The ADC needs to process only low frequencies, and such ADCs are low cost commodity items.  
      The invention may also be applied to intermediate frequency architectures. In such a case it will relax the LO suppression requirements of the filter(s). A lower suppression requirement allows fewer, smaller and cheaper filters. Alternatively, it allows the use of a lower intermediate frequency (IF). In the case of digital generation of the IF signal, a lower IF allows a cheaper digital-to-analog converter (DAC) to be used. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
      In the following, the invention will be described in greater detail by means of preferred embodiments and with reference to the attached drawings, in which  
       FIG. 1  shows one embodiment of an apparatus according to the invention;  
       FIG. 2  shows a timing diagram of the apparatus of  FIG. 1 ;  
       FIG. 3  shows another embodiment of an apparatus according to the invention;  
       FIG. 4  shows a timing diagram of the apparatus of  FIG. 3 ;  
       FIG. 5  shows still another embodiment of an apparatus according to the invention;  
       FIG. 6  shows a timing diagram relating to the apparatus of  FIG. 5 ;  
       FIG. 7  shows still another embodiment of an apparatus according to the invention;  
       FIG. 8  shows an embodiment of a method according to the invention. 
    
    
     DETAILED DESCRIPTION  
       FIG. 1  shows one embodiment of an apparatus of the invention. In short, the embodiment of  FIG. 1  shows a transmitter  100 , which receives a digital input signal. In the transmitter, digital compensation signals are formed to correct errors caused by components of the transmitter, and the digital compensation signals are used for modifying the digital input signal.  
      In the embodiment of  FIG. 1 , the functionality of the transmitter  100  has been split between a transmitting unit  110  and an observation receiver  140 . The transmitting unit includes functional entities, which together form a transmit signal to be transmitted on a radio path. The observation receiver  140  is a functional entity, which receives a portion of the transmit signal, observes possible errors in the transmit signal, and provides compensation signals to correct errors in the transmit signal.  
      In the transmitting unit  110 , the digital signal generator  112  provides an input signal, which may be either a baseband signal or a modulated intermediate frequency (IF) signal. Typically, baseband signals are in complex format including in-phase (I) and quadrature (Q) components. An intermediate frequency signal can be either in real or in complex format. The compensation signals are generally provided in complex format.  
      The digital signals are converted to the analog domain in digital-to-analog converters (DACs)  118  and  120 . If the digital signal is in real format, the lower DAC  120  is not used in the provision of the transmit signal, but it may still be needed in the cancellation loop when correcting errors in the transmitter. In the case of a digital baseband signal, the signal at the modulator  124  output becomes centered around a local oscillator frequency fLO. In the case of a real intermediate frequency signal at frequency fIF, the modulator  124  output contains signals at frequencies f LO +f IF  and f LO −f IF , one of which needs to be removed by filtering. If the IF signal is generated in complex format, both DACs  118 ,  120  are used in the signal path, and the quadrature modulator  124  functions as an image reject up-converter. Depending on the phasing between the I and Q signals, the output ideally shows either the frequency F LO +f IF  or F LO −f IF . In practice, the image is still present, but at a much lower power than the desired frequency, so that less filtering is needed to obtain its final rejection.  
      The RF signal at the modulator  124  output is further processed before it is fed to the antenna  132 . This processing typically includes several amplification stages  126 , power control (not shown), and filtering  130 . At some point in the chain, a sample of the signal is taken to the LO cancellation loop by a sampler  128 , which can be a coupler, for instance. The sampling point may in principle be anywhere between the modulator  124  output and the antenna. In some embodiments, the sampling point may be put as far downstream (close to the antenna  132 ) as possible to include as many LO leakage paths as possible, but before substantial amounts of power control and filtering which could interfere with the operation of the cancellation loop of the observation receiver  140 .  
      In the embodiment of  FIG. 1 , the sampled signal is fed via an RF switch  144  to a quadrature demodulator  146 . The local oscillator signal provided by the local oscillator  122  to the demodulator  146  is a copy of the oscillation signal fed to the modulator  124 , but delayed in a delay element  142  to correspond to the delay of the RF signal path leading to the demodulator  146 .  
      Important is the correct phasing of the oscillation signal. A leakage component detected at the I-output of the demodulator  146  may be reduced by providing a correction signal to the I-input of the modulator  124 . Similarly, a leakage component detected at the Q-output of the demodulator  146  may be reduced by providing a correction signal to the Q-input of the modulator  124 . However, simulations show that the phasing does not need to be very accurate. The phasing mainly effects loop dynamics, but not the final rejection. For the best performance, the phase error should be less than 30°. Even at phase errors between 45° and 90°, when the demodulated I signal correlates more to the transmitted Q than to the transmitted I signal, the loop still operates. However, the closer the phase error gets to 90°, the slower and the more oscillatory the settling becomes. It is not absolutely necessary to adjust the system to a demodulation phase error around 0°. With a phase error around 180°, the loop will operate correctly with inverted polarity. When the phase error is around 90° or 270°, the loop can be made to operate correctly by swapping the I- and Q-outputs of the observation receiver, possibly combined with polarity inversion.  
      As further shown by  FIG. 1 , at the demodulator  146  outputs the signal is directed to low-pass filters  148 ,  150  for filtering, in order to separate the detected LO leakage from the other signal components present in the RF signal. The low-pass filtered signals are sampled in analog-to-digital converters (ADC)  152 ,  154 , and multiplied in multipliers  156 ,  158  either by a number “a” or “b”, depending on the state of the RF switch  144 . The number “a” may be an exact or approximate inverse number of “b”. As an example, “a” can be (+1) and “b” can be (−1). In one embodiment, the sampling and multiplication interval is equal to the switch interval when one sample is taken at each switch interval.  
      Alternatively, more than one conversion sample may be taken per state of the RF switch, which is called oversampling. These samples are multiplied by a series of numbers. So if, for instance, [s 1   a , s 2   a , . . . , s 8   a ] are the samples produced in one switching interval, and [s 1   b , s 2   b , . . . , s 8   b ] are the samples produced in the other switching interval, the output of the multiplier  156  consists of the samples [a 1 ·s 1   a , a 2 ·s 2   a , . . . a 8 ·s 8   a , b 1 ·s 1   b , b 2 ·s 2   b , . . . b 8 ·s 8   b ]. One example of this kind of windowing is a rectangular window with a 1 =a 2 =. . . =a 8  (= 1 )and b 1 =b 2 =. . . =b 8  (=− 1 ).  
      Oversampling allows some of the low-pass filtering to be carried out in the digital domain, which might save costs in the implementation of the analog low-pass filters  148 ,  150 , but needs higher dynamic range in the ADCs  152 ,  154 . In the case of oversampling, it is also possible to multiply the conversion samples with a windowing function instead of a constant number as explained above. Windowing in the time domain is one possible way to achieve filtering in the frequency domain. The multiplier  156 ,  158  outputs are fed into loop filters  160 ,  162 , which are typically integrators. The loop filters average out the fast fluctuations at their input and determine the dynamic behaviour (e.g. settling time) of the loop. The loop filters can be either inverting or non-inverting. The correct polarity depends on the phasing of the signals in the loop. In  FIG. 1 , the control unit  164  controls that sampling in the ADCs  152 ,  154  and the multiplication in multipliers  156 ,  158  occur according to the control of the RF switch  144 . Finally, in the transmitter  100 , the loop filter outputs are digitally summed in summing units  114 ,  116  to the inputs of the DACs steering the quadrature modulator  124 .  
      The timings of the digital cancellation loop of the observation receiver  140  are shown in  FIG. 2 . The first graph  202  shows the timings of the switch  144  controlling the input of the radio frequency signal. The length of each period is designated as T. In the ON state, the radio frequency signal is passed, whereas the signal throughput is blocked in the OFF state of the switch.  
      Graph  204  shows the output of the demodulator  146 . For the sake of clarity, only the demodulated LO leakage component is shown and not the modulation on the transmitted signal. When the RF switch is in OFF state, the output of the demodulator is equal to its offset voltage V off . When the RF switch is in ON state, the detected LO leakage component ΔV LO  is added to the offset voltage and the output of the demodulator is V off +ΔV Lo .  
      The output of the low-pass filters  148 ,  150  is shown in graph  206 .  
      Sampling clock samples  208  provided by the sequencer  164  synchronously to the switch interval T are used in the analog-to-digital converters  152 ,  154  to determine the moments of conversion to the digital domain. The output  210  of the analog-to-digital converter  152 ,  154  is indicated by A corresponding to the offset voltage when the RF signal input is disabled (OFF state). The output  210  is indicated by B corresponding to the sum of the offset voltage and the leakage voltage when the RF signal input is enabled. The LO leakage voltage is thus the difference B−A.  
      The input to the integrator is inverted by using multiplication factors  212  provided by the multipliers  156 ,  158  to give output  214 . The output of the ADCs may be inverted (multiplication by −1) when the RF switch is OFF and the output is not inverted (multiplication by −1) when the RF switch is ON. Hence, the integrator input signal is given by  
         V   int     =     {               V   off     +     Δ   ⁢           ⁢     V   LO         ,         ON               -     V   off       ,         OFF               
 
      according to the states ON/OFF of the RF switch  144 . Assuming a 50% duty cycle, the offset voltage cancels in the averaging process in the loop filter, that is, in successive moments of time, voltages −A and A (A is included in B) are present in the loop filter input. Half of the detected LO leakage, which is present in B, remains, because the leakage is only passed in one of the two states of the RF switch.  FIG. 3  shows one embodiment of a transmitter  300  with leakage detection and compensation in the analog domain. The operation of the transmitter unit  310  is similar to the implementation of the transmitter unit  110  in  FIG. 1  except that the adding units  314 ,  316  come in the transmit chain after the digital-to-analog converters  318 ,  320 . That is, in the embodiment of  FIG. 3 , the compensation signals are added to an analog signal in contrast to the digital addition of compensation signals shown in  FIG. 1 .  
      In the observation receiver  340 , the difference as opposed to the observation receiver  110  in  FIG. 1 , starts at the output of the low-pass filters  348 ,  350  after the quadrature demodulator  346 . While in the digital implementation of  FIG. 1  the signals were first converted to the digital domain before polarity switching, in the analog implementation of  FIG. 3  the polarity of the analog signal may be switched directly after low-pass filtering. For differential signals the polarity can be switched without introducing new DC offset errors, just by swapping the inverted and non-inverted signal component. When the RF signal input is in OFF state, the baseband signal may be inverted, and correspondingly when the RF signal input is in ON state, the baseband signal is not inverted.  
      The output signals of the switches  356 ,  358  are fed to analog loop filters  360 ,  362 , typically integrators. Control unit  364  controls that the polarity switching of the switches  356 ,  358  is performed in alignment with the RF input enablement/disablement in the RF switch  344 .  
       FIG. 4  shows the timings in the analog embodiment of  FIG. 3 . The first graph  402  again shows the timings of the RF signal input, wherein T is the length of the ON/OFF state.  
      Graph  404  shows the differential output of demodulator  346 , i.e. the difference between its positive and negative output. Correspondingly to the timing in  FIG. 2 , in the OFF state of the RF switch, the offset voltage of the demodulator is obtained and in the ON state of the switch, the output is the sum of the offset voltage and the leakage voltage. Graph  406  shows the differential low-pass filter output. The base band switches  356 ,  358  may be controlled by the control shown by graph  408 , that is, at one RF switch interval T, the BB switch inverts the differential signal, and at another switch interval T, the BB switch does not invert the signal. This is highlighted by  FIG. 4  such that when the inversion takes place, signal portion C is inverted. At the moment when no inversion takes place, signal portion D is not inverted in graph  410 .  
      The control of the baseband switches should be somewhat delayed to the control of the RF switch, in order to accommodate the delay in the low-pass filters.  
       FIG. 5  shows still another embodiment of an observation receiver according to the invention. Only the I-branch is depicted in the Figure but the Q-branch may be implemented correspondingly. As shown in  FIG. 2 , the input of the loop filter alternates between the voltages A and B. Thus, the input signal contains an alternating current component at the switching frequency which appears at the output of the loop filter when its bandwidth is not sufficiently small. Alternatively, one may feed only the differences between pairs of samples B and A of low-pass filter outputs, which is achieved by the transmitter of  FIG. 5 .  
      In one of the states of the RF switch  544 , the ADC  552  outputs a set of samples [s 1   a , s 2   a , . . . , sna] the sample interval thus being a multiple of the switch interval. In the other state of the RF switch, the ADC  552  outputs a set of samples [s 1   b , s 2   b , . . . , snb]. Let the corresponding samples in the previous ON-OFF cycle of the RF switch be denoted by a prime, thus for instance s&#39; 1   b  is s 1   b  in the previous ON-OFF cycle. The samples are multiplied in a multiplication node  570 . The multiplication is carried out using sample-specific multiplication factors. Factors/series a=[a 1 , a 2 , . . . aN] are used for multiplying the samples A=[s 1   a , s 2   a , . . . , sna] and factors/series b=[b 1 , b 2 , . . . , bn] for multiplying the samples B=[s 1   b , s 2   b , . . . , snb]. The multiplier outputs are thus [s 1   a *a 1 , s 2   a *ba, . . . , sna*ba, s 1   b *b 1 , s 2   b *b  2 , . . . , snb*bn]. The multiplication series a and b may be substantially opposite to each other. Without oversampling, only one sample or multiplication factor exists per switch state, meaning n=1. With oversampling, n is more than 1. After the multiplier  570  the signal is branched. The upper branch is delayed by one switch interval of the RF switch. The samples at the output of the delay belong to one earlier state of the RF switch, so they are given by [s&#39; 1   b *b 1 , s&#39; 2   b *b 2 , . . . , s&#39;nb*bn, s 1   a *a 1 , s 2   a *ba, . . . , sna*ba]. The direct and delayed samples are summed, the result being the series [s 1   a *a 1 +s&#39; 1   b *b  1 , . . . , sna*an+s&#39;nb*bn, s 1   a *a 1 +s 1   b *b 1 , . . . , sna*an+snb*bn]. The effect of the delay in one branch is thus to time align the samples belonging to the ON state and OFF state of the RF switch, so that they can be combined simultaneously instead of alternately. This removes the switching frequency from the input of the loop filter.  
      As shown by  FIG. 5 , the generating means generates the compensation signal for a particular moment by using the filtered down-converted observation signals both in the ON and OFF states of the switching means.  
       FIG. 6  highlights alignment of sample windows. Sample sets A and B in graph  614 , each relating to one of the switch states, are aligned in time over each other. In this example sample set A is multiplied by −1 and sample set B by 1, so that the output signal is B−A shown by  616 .  
       FIG. 7  shows still another embodiment in the analog domain. During the time the RF switch  744  is open and the outputs of the low-pass filters  748 ,  750  have settled, the baseband switches marked by S 1   790 ,  794  are closed and the DC offset of the demodulator  746  is stored into the capacitors  780 ,  782 ,  784 ,  786 . This time is marked as “⇄1” in the output voltage  406  of the LPF in  FIG. 4 . During the time the RF switch  744  is closed and the out-puts of the low-pass filters  748 ,  750  have settled again, the baseband switches marked by S 2   791 ,  792 ,  795 ,  796  are closed while those marked by S 3   793 ,  797  are open. In  FIG. 4 , this time is marked as “⇄2, 3”. During this time, the demodulated LO leakage is passed to the loop filter, with the DC offset voltage stored in the capacitors subtracted from it. Outside the time interval “2, 3”, switches S 3  are closed and switches S 2  are open, so that the loop filter has no input signal and—assuming an integrating loop filter—its output will not change. The function of switch S 1  is thus to clamp the constant part of signal portion C of graph  406  in  FIG. 4  to zero and in that way to shift the whole graph in vertical direction. With the shown arrangement of switches, the circuit behaves similarly to that in the embodiment of  FIG. 5 , only passing the difference between the settled states of the LPFs  748 ,  750  to the loop filters. In one embodiment, the switches S 2  and S 3  are omitted and the signals to the loop filters are taken from the terminals of switches S 1 . In that way the whole signal portion D in graph  406  is passed to the loop filter, including the rising and falling slopes. As such these slopes are not harmful to loop operation, because they also contain some of the detected LO leakage. However, switches S 2  and S 3  are advantageous if the loop filter inputs draw DC bias currents. S 3  can short-circuit the loop filter inputs at moments when no full input signal is available, thus minimizing the offset errors due to the bias currents.  
       FIG. 8  shows one embodiment of a method according to the invention. A signal, either a baseband or an intermediate frequency signal, is generated and received  800  in a transmitter. This signal is up-converted  802  to a radio frequency signal. In one embodiment, an in-phase component and a quadrature component being 90 degrees phase-shifted to the in-phase component are provided. In another embodiment, only an in-phase component is provided.  
      The created RF signal is transmitted to the radio path in a usual manner, including certain filtering and amplifying steps. A portion from the created RF transmit signal is extracted and fed back  804  to an observation receiver of the transmitter. A copy of the oscillation signal used in up-conversion of the transmit signal is also fed to the observation receiver. The observation receiver includes an RF switch, which may toggle between having the RF signal input enabled and disabled. If the check  806  indicates that the RF signal input is enabled, the method proceeds to step  808 , whereas if the RF signal input is disabled, the method proceeds to step  810 .  
      Thus, a demodulator in the observation receiver receives the oscillation signal and the chopped RF signal. The output of the demodulator is filtered in a low-pass filter so as to reveal signal components close to the oscillation signal. In step  810 , a correction to the compensation signal only containing offset voltage as compared to ideal output, is formed from the filtered output of the demodulator. In step  808 , the output of the low-pass filters contains both the offset voltage and the leakage voltage, which is due to the leakage of the oscillation signal, the correction to the compensation signal being accordingly formed from both these signal components.  
      In step  811 , the formed corrections are used for adjusting the compensation signal used in modifying the input signal to the transmitter. Both corrections may be used sequentially, or they may be combined to provide a single adjustment. The adjustment may take place in the digital or in the analog domain. For adjustment in the digital domain, the analog signal needs to be converted to a digital signal, after which every other sample may be inverted before being used for modifying the compensation signal. For adjustment in the analog domain, the polarity of the low-pass filter output may be inverted before being used for modifying the compensation signal.  
      In step  812 , the formed compensation signal is used for modifying the input signal of the transmitter. The modification may take place in the digital or in the analog domain. If the formation of the compensation signal and the modification of the input signal do not take place in the same domain, conversion between those domains is required.  
      After the input signal has been modified, the RF signal is observed again to obtain a new adjustment to the compensation signal.  
      The figures above show only few embodiments of the invention. In another embodiment of the invention, a separate IQ modulator or vector modulator may be provided for the LO cancellation. In such a case the correction signals are passed to the separate modulator different from the modulator in the transmit path. The signals of the both modulators are then added to each other. This may be an alternative when the main up-converter is not DC coupled or does not have a quadrature input, like in intermediate frequency architectures without image rejection.  
      In still another embodiment of the invention, a digital loop may be arranged for providing digital compensation signals, but its outputs are converted to the analog domain via separate DACs such that the analog correction signals may be analogly added to the input signals. This embodiment may be applicable when the DACs in the transmit signal path are not DC-coupled to the quadrature modulator. Furthermore, the polarity inversion in the analog loop may also be realized in other ways than via switches. Further, the place of the periodic polarity inversion is not restricted to the places indicated in the figures, but can be at any place between the demodulator and the loop filter. In still another embodiment, after the DC offset removal, the processed outputs of the quadrature modulator may be converted to a single signal representing the power or amplitude of the LO leakage. Then, a search algorithm can be used for finding the proper combination of the I and Q components of the compensation signal such that the leakage vanishes. This embodiment provides the advantage that it allows arbitrary phase shifts between the LO inputs of the modulator and the demodulator.  
      The figures only show the necessary components for understanding the invention. For example, the following practical implementation aspects will be evident to those skilled in the art. Reconstruction filters may be needed at the DAC outputs, which filters may be either low-pass or band-pass ones depending on the signal and clock frequencies. Typically, the low-pass filtered outputs of the demodulator need some extra amplification before further processing. In the digital implementation, the extra amplification reduces the effect of quantization errors in the ADCs, and in the analog implementation it reduces the effect of DC offsets in the loop filters. Some of the DC offset in the demodulator is caused by LO leakage in the demodulator that is reflected back from its RF input. Therefore it is assumed that the impedance seen by the RF input of the demodulator does not depend on the state of the RF switch. This can be realized by using a non-reflective switch and/or buffer amplifiers and/or attenuators.  
      The invention may be implemented in hardware by using the disclosed or corresponding components.  
      It will be obvious to a person skilled in the art that as the technology advances, the inventive concept can be implemented in various ways. The invention and its embodiments are not limited to the examples described above but may vary within the scope of the claims.