Patent Publication Number: US-7719351-B2

Title: Autozeroing current feedback instrumentation amplifier

Description:
REFERENCE TO CO-PENDING APPLICATIONS FOR PATENT 
   The present Application for Patent is related to co-pending U.S. patent application Ser. No. 11/804,490, now U.S. Pat. No. 7,573,327, entitled “AUTOZEROING CURRENT FEEDBACK INSTRUMENTATION AMPLIFIER”, by Michiel Pertijs et al., filed May 17, 2007, assigned to the assignee hereof, and expressly incorporated by reference herein. 
   BACKGROUND 
   1. Field 
   Embodiments generally relate to the field of current feedback instrumentation amplifiers. 
   2. Background 
   Instrumentation amplifiers are commonly used to amplify small differential input voltages while rejecting common-mode input voltages. A desired feature of such amplifiers is a low input-referred offset voltage combined with a low input current. The latter can be achieved by using a MOS input stage, but such an input stage typically results in a high offset voltage. 
   Another desired feature of instrumentation amplifiers is that their input range includes the negative supply rail, so that they can be connected to a grounded signal source in a single-supply system. This is not possible with a conventional 3-opamp instrumentation amplifier topology. 
   This limitation to some extent has been overcome by using a current-feedback topology with PMOS input transistors. The PMOS transistors transfer the differential input voltage to a resistor connected between their sources, resulting in a current proportional to the differential input voltage. The PMOS transistors at the same time provide the required common-mode level-shift to be able to make this voltage-to-current conversion with an input voltage at ground level. In the rest of the amplifier, the generated current is converted back into an output voltage using a second resistor. 
     FIG. 1  shows a block diagram of one conventional current-feedback instrumentation amplifier  100 . The differential input voltage V in  is converted to a current by transconductance amplifier g 2 . As described above, amplifier g 2  has PMOS input transistors which enable it to sense input signals at the negative supply rail. The difference between the output voltage V out  and a reference voltage V ref  is scaled down by a resistive divider, consisting of R 1  and R 2 , to provide a feedback voltage V fb . This is applied to a second transconductance amplifier g 3 . The feedback loop, closed by the output stage g 1 , ensures that the output current of g 3  equals that of g 2 . It is appreciated that the output stage, here shown as a single Miller-compensated transconductance stage, can in practice consist of multiple stages. If the two transconductances g 2  and g 3  are equal, V fb  equals V in , and therefore the output voltage equals:
   V   out   =V   ref +( R   1   +R   2 )/ R   2   ·V   in   (1) 
   In the more general case that the two transconductances are not equal, the output voltage equals:
 
 V   out   =V   ref   +g   2   /g   3 ·( R   1   +R   2 )/ R   2   ·V   in   (2)
 
   In addition to its ability to sense input voltages at the negative supply rail, amplifier  100  has the attractive feature that its output can swing rail-to-rail, which is important in low-voltage applications. 
   However, circuit  100  is disadvantageous in that the offsets of transconductance amplifiers g 2  and g 3  add directly to the input voltage, and therefore need to be compensated for.  FIG. 2  illustrates one conventional amplifier  200  that employs chopper switches  210  and  220  added at the input of g 2  and g 3  to periodically reverse the polarity of the input and feedback signal. An additional chopper switch  230  at the input of the output stage restores the original polarity. This configuration effectively modulates the offset of the transconductance amplifiers to the chopper frequency, where it can, in principle, be filtered out. 
   An important disadvantage of using chopping to eliminate the offset in current-feedback instrumentation amplifiers is that the modulated offset results in spurious AC signals at the output of the amplifier  200 . For example, the output of amplifier  200  may actually appear as a sawtooth signal. Since the output of an instrumentation amplifier is typically sampled by an analog-to-digital converter, such spurious signal may result in measurement errors unless they are filtered out. Conventional implementations have attempted to reduce and filter these spurious signals by using a continuous (non-chopped) feedforward path and various extra offset-compensation loops. This, however, leads to a very large and complex system. 
   Another important disadvantage of using chopping is that the input source is exposed to a switched capacitive load consisting of the input capacitance C in2  of transconductance amplifier g 2 . Due to the periodic polarity reversal, this input capacitance has to be alternately charged to +V in  and −V in . The associated current results in an input offset current. Effectively, this reduces the input impedance of the instrumentation amplifier (e.g., amplifier  200 ) to:
 
 R   in =1/(2· f   chop   ·C   in2 ).  (3)
 
For typical values of f chop =10 kHz and C in2 =1 pF, the input impedance is 50 MΩ. In contrast, non-chopped instrumentations amplifiers with MOS inputs typically achieve input impedances on the order of 10 GΩ. This reduced impedance due to chopping can cause significant gain errors when reading out a high-impedance signal source. A similar problem occurs at the input of transconductance amplifier g 3 , whose input capacitance C in3  presents a switched load to the feedback network.
 
   Thus, conventional current feedback instrumentation amplifiers do not provide a simple way to reduce input offsets while at the same time maintaining high input impedance and avoiding spurious signals at the output. 
   SUMMARY 
   This summary is provided to introduce a selection of concepts in a simplified form that are further described below in the Detailed Description. This summary is not intended to identify key features or essential features of the claimed subject matter, nor is it intended to be used to limit the scope of the claimed subject matter. 
   An embodiment is directed to an instrumentation amplifier. The instrumentation amplifier includes an output stage for generating an output voltage, a low-frequency path coupled with the output stage, and a high-frequency path coupled with the output stage. The high-frequency path dominates the low-frequency path at frequencies above a particular frequency, and the low-frequency path dominates the high-frequency path at frequencies below the particular frequency. The low-frequency path includes an input stage for sensing a differential input and generating an intermediate current based thereon, a feedback stage coupled with the input and output stages, the feedback stage for generating a feedback current based on the output voltage, and an auto-zeroing circuit coupled with the input, feedback, and output stages, the auto-zeroing circuit for generating a nulling current. The nulling current compensates for errors in the intermediate and feedback currents resulting from input offsets in the input and feedback stages. 
   Thus, embodiments provide technology allowing for instrumentation amplifiers with very low input-referred offset, low input current, and low level spurious switching signals at the output. Additionally, spurious signals may be further reduced by adding a high-frequency feedforward path. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings, which are incorporated in and form a part of this specification, illustrate embodiments of the invention and, together with the description, serve to explain the principles of embodiments of the invention: 
       FIG. 1  shows a block diagram of one conventional current-feedback instrumentation amplifier. 
       FIG. 2  illustrates one conventional amplifier that employs chopper switches added at the input and output of g 2  and g 3  from  FIG. 1  to modulate the offset of g 2  and g 3  away from DC. 
       FIG. 3  illustrates a block diagram of a current feedback amplifier, in accordance with various embodiments of the present invention. 
       FIG. 4  illustrates a schematic of a current feedback amplifier, in accordance with various embodiments of the present invention. 
       FIG. 5  illustrates a block diagram of a current feedback amplifier, including a high-frequency feedforward path, in accordance with various embodiments of the present invention. 
       FIG. 6  illustrates a schematic of a current feedback instrumentation amplifier, including a high-frequency feedforward path, in accordance with various embodiments of the present invention. 
       FIG. 7  illustrates a schematic of a current feedback instrumentation amplifier, including parallel input stages, in accordance with various embodiments of the present invention. 
       FIG. 8  illustrates a schematic of a current feedback instrumentation amplifier, including parallel input stages, in accordance with various embodiments of the present invention. 
       FIG. 9  illustrates a block diagram of a current feedback instrumentation amplifier, including parallel input stages and a high-frequency feedforward path, in accordance with various embodiments of the present invention. 
       FIG. 10  illustrates a schematic of a current feedback instrumentation amplifier, including parallel input stages and a high-frequency feedforward path, in accordance with various embodiments of the present invention. 
       FIG. 11  illustrates a schematic of a current feedback instrumentation amplifier, including a current buffer stage, in accordance with various embodiments of the present invention. 
       FIG. 12  illustrates an input stage of an instrumentation amplifier that includes pre-charging circuitry, in accordance with various embodiments of the present invention. 
       FIG. 13  illustrates a flowchart of a process for reducing effects of offsets in a current feedback instrumentation amplifier, in accordance with various embodiments of the present invention. 
       FIG. 14  illustrates a flowchart of a process for generating a nulling current, in accordance with various embodiments of the present invention. 
       FIG. 15  illustrates a flowchart of a process for switching an instrumentation amplifier from an amplification configuration to an auto-zero configuration, in accordance with various embodiments of the present invention. 
       FIGS. 16A-16B  illustrate a flowchart for a process for reducing the effects of offsets in an instrumentation amplifier, in accordance with various embodiments of the present invention. 
       FIG. 17  illustrates a flowchart for a process of calibrating a nulling current, in accordance with various embodiments of the present invention. 
   

   DETAILED DESCRIPTION 
   Reference will now be made in detail to the preferred embodiments of the invention, examples of which are illustrated in the accompanying drawings. While the invention will be described in conjunction with the preferred embodiments, it will be understood that they are not intended to limit the invention to these embodiments. On the contrary, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the claims. Furthermore, in the detailed description of the present invention, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, it will be obvious to one of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well known methods, procedures, components, and circuits have not been described in detail as not to unnecessarily obscure aspects of the present invention. 
   Overview 
   Generally speaking, embodiments provide technology for reducing input offsets in current feedback instrumentation amplifiers. The technology involves using auto-zeroing circuitry to null an offset of an input stage. In one embodiment, this is achieved by periodically switching-in the auto-zeroing circuitry. As a result, embodiments are able to achieve very low input-referred offset, low input current, and low-level spurious switching signals at the output. Additionally, spurious signals may be further reduced by adding a high-frequency feedforward path. 
   Exemplary Circuits, in Accordance with Various Embodiments 
     FIG. 3  illustrates a block diagram of a current feedback amplifier  300 , in accordance with various embodiments of the present invention. Amplifier  300  includes an input stage  320 , an output stage  310 , a feedback stage  330 , and a feedback network  350 . The feedback network is operable to generate a feedback voltage V fb  from the output voltage V out  and the reference voltage V ref  and thus defines the gain of the amplifier. The amplifier  300  also advantageously includes an auto-zero circuit  340  coupled to the input stage  320 , the output stage  310 , and the feedback stage  330 . The auto-zero circuit  340  is operable to switch the amplifier  300  between an amplification configuration corresponding to an amplification phase and an auto-zeroing configuration corresponding to an auto-zeroing phase. During the amplification phase, the amplifier  300  is operable to perform normal amplification operations. During the auto-zeroing phase, the auto-zero circuit  340  is operable to null offset currents generated by the input stage  320  and the feedback stage  330 . 
   In one embodiment, the auto-zero circuit  340  nulls the offset currents by shorting inputs of the input stage  320  and the feedback stage  330  to respective common mode voltages. Subsequently, the auto-zero circuit  340  may then measure corresponding offset currents generated by the input stage  320  and the feedback stage  330  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the input stage  320  and the feedback stage  330 . 
   When the auto-zero circuit  340  switches the amplifier  300  back to the amplification configuration, the auto-zero circuit  340  continues to generate the nulling current, thereby reducing or even eliminating offsets in the amplifier  300 . In one embodiment, the auto-zero circuit  340  periodically switches between amplification and auto-zeroing configurations so as to periodically recalibrate the nulling current. 
   It should be appreciated that amplifier  300  may be achieved in a number of ways. For example,  FIG. 4  illustrates a schematic of a current feedback amplifier  400 , in accordance with various embodiments of the present invention. In amplifier  400 , tranconductance amplifier  411  serves as an output stage, such as output stage  310  in amplifier  300 , tranconductance amplifier  412  serves as an input stage, such as input stage  320  of amplifier  300 , tranconductance amplifier  413  serves as a feedback stage, such as feedback stage  330  of amplifier  300 , and resistors  461  and  462  serve as feedback network, such as feedback network  350  of amplifier  300 . Although embodiments may be described herein with reference to single tranconductance amplifiers, it should be appreciated that embodiments are not limited as such. For example, a cascade of multiple stages with appropriate frequency compensation may be used in place of the single tranconductance amplifier  411  in order to obtain a higher gain. Capacitors  451  and  452  serve as frequency compensators for the tranconductance amplifier  411 , thus forming a Miller-compensated output stage. Resistors  461  and  462 , together with the reference voltage V ref , generate the feedback voltage V fb  based on the output voltage V out . V fb  is fed back as an input to the feedback tranconductance amplifier  413 . 
   Switches  431 - 436  and  441 - 446 , tranconductance amplifiers  414  and  415 , and capacitors  453  and  454  function together as an auto-zero circuit, such as auto-zero circuit  340  of amplifier  300 . It should be appreciated that switches  431 - 436  and  441 - 446  may be any of a number of devices capable of performing a switching function. In one embodiment, the switches  431 - 436  and  441 - 446  serve to switch the amplifier  400  between amplification and auto-zeroing configurations. For example, an amplification configuration may correspond to switches  441 - 446  being closed and switches  431 - 436  being open. Conversely, an auto-zeroing configuration may correspond to switches  431 - 436  being closed and switches  441 - 446  being open. 
   During the auto-zeroing phase, the inputs of the tranconductance amplifiers  412  and  413  are respectively shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb . Any input offsets of amplifiers  412  and  413  cause an offset current that flows into the integrator formed by tranconductance amplifier  414  and capacitors  453  and  454 . The output of this integrator then drives the tranconductance amplifier  415  to generate a nulling current, which effectively nulls the offset current. 
   At the end of the auto-zeroing phase, switches  431 - 436  open. As a result, the voltage at the output of the integrator around amplifier  414  is held so that amplifier  415  continues nulling the offset current at the outputs of amplifiers  412  and  413 . 
   Subsequently, in the amplification phase, switches  441 - 446  are closed. V in  and V fb  are applied to amplifiers  412  and  413 , respectively, and the summed output current of amplifiers  412 ,  413 , and  415  is coupled with the output stage (i.e., amplifier  411 , etc.). The amplifier  400  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by amplifier  415  ensures that the input-referred offset voltages of amplifiers  412 - 413  do not contribute to the output voltage. Thereafter, in a subsequent auto-zeroing phase, the Miller-compensated output stage formed by amplifier  411  and capacitors  451 - 452  holds the output voltage while amplifiers  412 - 413  are auto-zeroed again. 
   In some instances, the gating of the input signal may result in detection of components of the input signal (including noise) at harmonics of the clock frequency. Such components may mix with the clock signal and be modulated down to baseband. Consequently, this may result in errors and increased noise at the output of amplifiers  300  and  400 . 
   In one embodiment, this mixing may be prevented by using a high-frequency feedforward path.  FIG. 5  illustrates a block diagram of a current feedback amplifier  500 , including a high-frequency feedforward path, in accordance with various embodiments of the present invention. Amplifier  500  includes input stages  520  and  570 , an output stage  510 , feedback stages  530  and  580 , and a feedback network  550 . The amplifier  500  also advantageously includes an auto-zero circuit  540  coupled to the input stage  520 , the output stage  510 , and the feedback stage  530 . The auto-zero circuit  540  is operable to switch the amplifier  500  between an amplification configuration corresponding to an amplification phase and an auto-zeroing configuration corresponding to an auto-zeroing phase. During the amplification phase, the amplifier  500  is operable to perform normal amplification operations. During the auto-zeroing phase, the auto-zero circuit  540  is operable to null offset currents generated by the input stage  520  and the feedback stage  530 . 
   In one embodiment, the auto-zero circuit  540  nulls the offset currents by shorting inputs of the input stage  520  and the feedback stage  530  to respective common mode voltages. Subsequently, the auto-zero circuit  540  may then measure corresponding offset currents generated by the input stage  520  and the feedback stage  530  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the input stage  520  and the feedback stage  530 . 
   When the auto-zero circuit  540  switches the amplifier  500  back to the amplification configuration, the auto-zero circuit  540  continues to generate the nulling current, thereby reducing or even eliminating offsets in the amplifier  500 . In one embodiment, the auto-zero circuit  540  periodically switches between amplification and auto-zeroing configurations so as to periodically recalibrate the nulling current. 
   For low frequencies (e.g., below the clock frequency), the auto-zero path comprising input stage  520 , feedback stage  530 , and auto-zero circuit  540  is dominant, and the amplifier  500  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by the auto-zero circuit  540  ensures that the input-referred offsets of input stage  520  and feedback stage  530  do not contribute to the output voltage. Thereafter, in a subsequent auto-zeroing phase, the output stage  510  may hold the output voltage while the input stage  520  and the feedback stage  530  are auto-zeroed again. 
   At high frequencies, the feedforward path comprising input stage  570  and feedback stage  580  is dominant. Above a threshold frequency, the feedforward path ensures that the feedback signal V fb  can track the input signal V in . As a result, even if mixing occurs due to the gating at the inputs of input stage  520  and feedback stage  530 , the resulting mixing products cancel. 
   An additional advantage of the feedforward path is that it attenuates switching transients produced by the auto-zeroed input stage. The lower the threshold frequency, the higher the relative gain of the feedforward path at the clock frequency and its harmonics, and therefore the better the attenuation of such switching transients. 
   It is appreciated that amplifier  500  may be achieved in a number of ways. For example,  FIG. 6  illustrates a schematic of a current feedback instrumentation amplifier  600 , including a high-frequency feedforward path, in accordance with various embodiments of the present invention. In amplifier  600 , tranconductance amplifiers  611  and  616  together serve as an output stage, such as output stage  510  of amplifier  500 , tranconductance amplifiers  612  and  617  serve as input stages, such as input stages  520  and  570  of amplifier  500 , tranconductance amplifiers  613  and  618  serve as feedback stages, such as feedback stages  530  and  580  of amplifier  500 , and resistors  661  and  662  together serve as feedback network, such as feedback network  550  of amplifier  500 . Capacitors  651  and  652  serve as frequency compensators for the tranconductance amplifier  611 , thus forming a nested-Miller-compensated output driver stage. Additionally, tranconductance amplifier  616 , along with capacitors  656 - 657 , serves as a Miller-compensated intermediate stage to amplifier  600 . Resistors  661  and  662  together with the reference voltage V ref  generate the feedback voltage V fb  based on the output voltage V out . V fb  is fed back as an input to the feedback tranconductance amplifiers  613  and  618 . 
   Switches  631 - 636  and  641 - 646 , tranconductance amplifiers  614  and  615 , and capacitors  653  and  654  function together as an auto-zero circuit, such as auto-zero circuit  340  of amplifier  300 . It should be appreciated that switches  631 - 636  and  641 - 646  may be any of a number of devices capable of performing a switching function. In one embodiment, the switches  631 - 636  and  641 - 646  serve to switch the amplifier  600  between amplification and auto-zeroing configurations. For example, an amplification configuration may correspond to switches  641 - 646  being closed and switches  631 - 636  being open. Conversely, an auto-zeroing configuration may correspond to switches  631 - 636  being closed and switches  641 - 646  being open. 
   During the auto-zeroing phase, the inputs of the tranconductance amplifiers  612  and  613  are respectively shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb . Any input offsets of amplifiers  612  and  613  causes an offset current that flows into the integrator formed by tranconductance amplifier  614  and capacitors  653  and  654 . The output of this integrator then drives the tranconductance amplifier  615  to generate a nulling current, which effectively nulls the offset current. 
   At the end of the auto-zeroing phase, switches  631 - 636  open. As a result, the voltage at the output of the integrator around amplifier  614  is held so that amplifier  615  continues nulling the offset current at the outputs of amplifiers  612  and  613 . Subsequently, in the amplification phase, switches  641 - 646  are closed. V in  and V fb  are applied to amplifiers  612  and  613 , respectively, and the summed output current of amplifiers  612 ,  613 , and  615  is coupled with the intermediate stage (i.e., amplifier  616 ). 
   For low frequencies (e.g., below the clock frequency), the auto-zero path comprising amplifiers  611 - 616  is dominant, and the amplifier  600  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by amplifier  615  ensures that the input-referred offset voltages of amplifiers  612 - 613  do not contribute to the output voltage. Thereafter, in a subsequent auto-zeroing phase, the nested-Miller-compensated output stage formed by amplifier  611  and capacitors  651 - 652  and amplifier  616  and capacitors  656 - 657  hold the output voltage while amplifiers  612 - 613  are auto-zeroed again. 
   At high frequencies, the feedforward path comprising amplifiers  617 - 618  is dominant. The feedforward path, together with the output amplifier  611 , forms a regular Miller-compensated two-stage amplifier with approximately 20 dB/dec roll-off. This type of frequency compensation is known as “multi-path nested-Miller compensation” and has been used in conventional op-amps, but without application to auto-zeroed instrumentation amplifiers. 
   In one embodiment, the frequency at which the feedforward path starts to dominate is:
 
ω pz   =g   618   /C   651 ,  (1)
 
(assuming C 651 =C 652  and g 617 =g 618 ). In a preferred embodiment, this frequency is chosen to be below the clock frequency. Above ω pz , the feedforward path ensures that the feedback signal V fb  can track the input signal V in . As a result, even if mixing occurs due to the gating at the inputs of amplifiers  612 - 613 , the resulting mixing products cancel.
 
   The above-referenced mixing problems may alternatively be solved by using a dual-input-stage “ping-pong” architecture.  FIG. 7  illustrates a schematic of a current feedback instrumentation amplifier  700 , including parallel input stages, in accordance with various embodiments of the present invention. Amplifier  700  includes first and second input stages  720  and  725 , an output stage  710 , first and second feedback stages  730  and  735 , and a feedback network  750 . The amplifier  700  also advantageously includes a first auto-zero circuit  740  coupled to the input stage  720 , the output stage  710 , and the feedback stage  730 . The amplifier  700  further includes a second auto-zero circuit  745  coupled to the input stage  725 , the output stage  710 , and the feedback stage  735 . 
   In one embodiment, the auto-zero circuits  740  and  745  serve to switch the amplifier  700  between first and second configurations corresponding to first and second phases of operation. For example, the first configuration may correspond to an auto-zero configuration of the auto-zero circuit  740  and an amplification configuration of the auto-zero circuit  745 . Conversely, a second configuration may correspond to an auto-zero configuration of the auto-zero circuit  745  and an amplification configuration of the auto-zero circuit  740 . 
   During the first phase, the first input stage  720  and the first feedback stage  730  are auto-zeroed while the second input stage  725  and the second feedback stage  735  perform the amplification functions of amplifier  700 . Conversely, during the second phase, the second input stage  725  and the second feedback stage  735  are auto-zeroed while the first input stage  720  and the first feedback stage  730  perform the amplification functions of amplifier  700 . 
   Thus, during the first phase, the auto-zero circuit  740  is operable to null offset currents generated by the first input stage  720  and the first feedback stage  730 . In one embodiment, the auto-zero circuit  740  nulls the offset currents by shorting inputs of the first input stage  720  and the first feedback stage  730  to respective common mode voltages. Subsequently, the auto-zero circuit  740  may then measure corresponding offset currents generated by the first input stage  720  and the first feedback stage  730  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the first input stage  720  and the first feedback stage  730 . 
   Concurrently, V in  is applied to the second input stage  725 , V fb  is applied to the second feedback stage  735 , and the second input stage  725  and second feedback stage  735  are coupled with the output stage  710  via the second auto-zero circuit  745 . The amplifier  700  then operates similar to a traditional current feedback instrumentation amplifier, except that a nulling current injected by the second auto-zero circuit  745  (which is calibrated in the second phase, discussed below) ensures that the input-referred offset voltages of the second input stage  725  and the second feedback stage  735  do not contribute to the output voltage. 
   At the end of the first phase, the first auto-zero circuit  740  changes from an auto-zero configuration to an amplification configuration, and the second auto-zero circuit  745  changes from an amplification configuration to an auto-zero configuration. Thereafter, the auto-zero circuit  740  continues nulling the offset current at the outputs of the first input stage  720  and the first feedback stage  730 . 
   Subsequently, in the second phase, V in  is applied to the first input stage  720 , V fb  is applied to the first feedback stage  730 , and the first input stage  720  and the first feedback stage  730  are coupled with the output stage  710  via the first auto-zero circuit  740 . The amplifier  700  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by the first auto-zero circuit  740  ensures that the input-referred offset voltages of the first input stage  720  and the first feedback stage  730  do not contribute to the output voltage. 
   During the second phase, while the first input stage  720  and the first feedback stage  730  are performing amplification functions, the second auto-zero circuit  745  is operable to null offset currents generated by the second input stage  725  and the second feedback stage  735 . In one embodiment, the second auto-zero circuit  745  nulls the offset currents by shorting inputs of the second input stage  725  and the second feedback stage  735  to respective common mode voltages. Subsequently, the second auto-zero circuit  745  may then measure corresponding offset currents generated by the second input stage  725  and the second feedback stage  735  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the second input stage  725  and the second feedback stage  735 . 
   During operation, the auto-zeroing circuits  740  and  745  of amplifier  700  periodically switch amplifier  700  between the first configuration and the second configuration, ensuring that the input stages  720  and  725  and feedback stages  730  and  735  are periodically recalibrated. Thus, this “ping-pong” operation ensures that there is continuously an offset-free stage in the signal path. 
   It is appreciated that amplifier  700  may be achieved in a number of ways. For example,  FIG. 8  illustrates a schematic of a current feedback instrumentation amplifier  800 , including parallel input stages, in accordance with various embodiments of the present invention. In amplifier  800 , tranconductance amplifier  811  serves as an output stage, such as output stage  710  in amplifier  700 , tranconductance amplifiers  812  and  822  serve as first and second input stages, such as input stages  720  and  725  of amplifier  700 , tranconductance amplifiers  813  and  823  serve as first and second feedback stages, such as feedback stages  730  and  735  of amplifier  700 , and resistors  861  and  862  together serve as a feedback network, such as feedback network  750  of amplifier  700 . Although embodiments may be described herein with reference to single tranconductance amplifiers, it should be appreciated that embodiments are not limited as such. For example, a cascade of multiple stages with appropriate frequency compensation may be used in place of the single tranconductance amplifier  811  in order to obtain a higher gain. Capacitors  851  and  852  serve as frequency compensators for the tranconductance amplifier  811 , thus forming a Miller-compensated output stage. Resistors  861  and  862 , together with the reference voltage V ref , generate the feedback voltage V fb  based on the output voltage V out . V fb  is fed back as an input to the feedback tranconductance amplifiers  813  and  823 . 
   Switches  831 - 836  and  841 - 846 , tranconductance amplifiers  814  and  815 , and capacitors  853  and  854  function together as a first auto-zero circuit, such as auto-zero circuit  740  of amplifier  700 . Similarly, switches  871 - 876  and  881 - 886 , tranconductance amplifiers  824  and  825 , and capacitors  858  and  859  function together as a second auto-zero circuit, such as auto-zero circuit  745  of amplifier  700 . It should be appreciated that switches  831 - 836 ,  841 - 846 ,  871 - 876 , and  881 - 886  may be any of a number of devices capable of performing a switching function. In one embodiment, the switches  831 - 836 ,  841 - 846 ,  871 - 876 , and  881 - 886  serve to switch the amplifier  800  between first and second configurations corresponding to first and second phases of operation. For example, the first configuration may correspond to switches  831 - 836  and  871 - 876  being closed and switches  841 - 846  and  881 - 886  being open. Conversely, a second configuration may correspond to switches  841 - 846  and  881 - 886  being closed and switches  831 - 836  and  871 - 876  being open. 
   During the first phase, the first input stage and the first feedback stage are auto-zeroed while the second input stage and the second feedback stage perform the amplification functions of amplifier  800 . Conversely, during the second phase, the second input stage and the second feedback stage are auto-zeroed while the first input stage and the first feedback stage perform the amplification functions of amplifier  800 . 
   Thus, during the first phase, the inputs of the tranconductance amplifiers  812  and  813  are shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb , respectively. Any input offsets of amplifiers  812  and  813  cause an offset current that flows into the integrator formed by tranconductance amplifier  814  and capacitors  853  and  854 . The output of this integrator then drives the tranconductance amplifier  815  to generate a nulling current, which effectively nulls the offset current. 
   Concurrently, V in  and V fb  are applied to amplifiers  822  and  823 , respectively, and the summed output current of amplifiers  822 ,  823 , and  825  is coupled with the output stage (i.e., amplifier  811 , etc.). The amplifier  800  then operates similar to a traditional current feedback instrumentation amplifier, except that a nulling current injected by amplifier  825  (which is calibrated in the second phase, discussed below) ensures that the input-referred offset voltages of amplifiers  822 - 823  do not contribute to the output voltage. 
   At the end of the first phase, switches  831 - 836  and  871 - 876  open. As a result, the voltage at the output of the integrator around amplifier  814  is held so that amplifier  815  continues nulling the offset current at the outputs of amplifiers  812  and  813 . 
   Subsequently, in the second phase, switches  841 - 846  and  881 - 886  are closed. V in  and V fb  are applied to amplifiers  812  and  813 , respectively, and the summed current of amplifiers  812 ,  813 , and  815  is coupled with the output stage (i.e., amplifier  811 , etc.). The amplifier  800  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by amplifier  815  insures that the input-referred offset voltages of amplifiers  812 - 813  do not contribute to the output voltage. 
   During the second phase, while the first input stage and the first feedback stage are performing amplification functions, the second input stage and the second feedback stage are auto-zeroed. In other words, the inputs of the tranconductance amplifiers  822  and  823  are shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb , respectively. Any input offsets of amplifiers  822  and  823  cause an offset current that flows into the integrator formed by tranconductance amplifier  824  and capacitors  858  and  859 . The output of this integrator then drives the tranconductance amplifier  825  to generate a nulling current, which effectively nulls the offset current. 
   During operation, the auto-zeroing circuits of amplifier  800  periodically switch amplifier  800  between the first configuration and the second configuration, ensuring that the input stages and feedback stages are periodically recalibrated. Thus, this “ping-pong” operation ensures that there is continuously an offset-free stage in the signal path. 
   It is appreciated that switching between parallel input and feedback stages may cause corresponding transients to appear in the output signal. Therefore, in one embodiment, a high-frequency feedforward path may be used in combination with the ping-pong architecture of  FIG. 8 .  FIG. 9  illustrates a block diagram of a current feedback instrumentation amplifier  900 , including parallel input stages and a high-frequency feedforward path, in accordance with various embodiments of the present invention. Amplifier  900  includes first, second, and third input stages  920   925 , and  970 , an output stage  910 , first, second, and third feedback stages  930 ,  935 , and  980 , and a feedback network  950 . The amplifier  900  also advantageously includes a first auto-zero circuit  940  coupled to the input stage  920 , the output stage  910 , and the feedback stage  930 . The amplifier  900  further includes a second auto-zero circuit  945  coupled to the input stage  925 , the output stage  910 , and the feedback stage  935 . 
   In one embodiment, the auto-zero circuits  940  and  945  serve to switch the amplifier  900  between first and second configurations corresponding to first and second phases of operation. For example, the first configuration may correspond to an auto-zero configuration of the auto-zero circuit  940  and an amplification configuration of the auto-zero circuit  945 . Conversely, a second configuration may correspond to an auto-zero configuration of the auto-zero circuit  945  and an amplification configuration of the auto-zero circuit  940 . 
   During the first phase, the first input stage  920  and the first feedback stage  930  are auto-zeroed while the second input stage  925  and the second feedback stage  935  perform the amplification functions of amplifier  900 . Conversely, during the second phase, the second input stage  925  and the second feedback stage  935  are auto-zeroed while the first input stage  920  and the first feedback stage  930  perform the amplification functions of amplifier  900 . 
   Thus, during the first phase, the auto-zero circuit  940  is operable to null offset currents generated by the first input stage  920  and the first feedback stage  930 . In one embodiment, the auto-zero circuit  940  nulls the offset currents by shorting inputs of the first input stage  920  and the first feedback stage  930  to respective common mode voltages. Subsequently, the auto-zero circuit  940  may then measure corresponding offset currents generated by the first input stage  920  and the first feedback stage  930  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the first input stage  920  and the first feedback stage  930 . 
   Concurrently, V in  is applied to the second input stage  925 , and V fb  is applied to the second feedback stage  935 , and the second input stage  925  and second feedback stage  935  are coupled with the output stage  910  via the second auto-zero circuit  945 . The amplifier  900  then operates similar to a traditional current feedback instrumentation amplifier, except that a nulling current injected by the second auto-zero circuit  945  (which is calibrated in the second phase, discussed below) ensures that the input-referred offset voltages of the second input stage  925  and the second feedback stage  935  do not contribute to the output voltage. 
   At the end of the first phase, the first auto-zero circuit  940  changes from an auto-zero configuration to an amplification configuration, and the second auto-zero circuit  945  changes from an amplification configuration to an auto-zero configuration. Thereafter, the auto-zero circuit  940  continues nulling the offset current at the outputs of the first input stage  920  and the first feedback stage  930 . 
   Subsequently, in the second phase, V in  is applied to the first input stage  920 , V fb  is applied to the first feedback stage  930 , and the first input stage  920  and the first feedback stage  930  are coupled with the output stage  910  via the first auto-zero circuit  940 . The amplifier  900  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by the first auto-zero circuit  940  ensures that the input-referred offset voltages of the first input stage  920  and the first feedback stage  930  do not contribute to the output voltage. 
   During the second phase, while the first input stage  920  and the first feedback stage  930  are performing amplification functions, the second auto-zero circuit  945  is operable to null offset currents generated by the second input stage  925  and the second feedback stage  935 . In one embodiment, the second auto-zero circuit  945  nulls the offset currents by shorting inputs of the second input stage  925  and the second feedback stage  935  to respective common mode voltages. Subsequently, the second auto-zero circuit  945  may then measure corresponding offset currents generated by the second input stage  925  and the second feedback stage  935  and generate a nulling current based thereon. The nulling current serves to compensate for the offset currents generated by the second input stage  925  and the second feedback stage  935 . 
   During operation, the auto-zeroing circuits  940  and  945  of amplifier  900  periodically switch amplifier  900  between the first configuration and the second configuration, ensuring that the input stages  920  and  925  and feedback stages  930  and  935  are periodically recalibrated. Thus, this “ping-pong” operation ensures that there is continuously an offset-free stage in the signal path. 
   For low frequencies (e.g., below the clock frequency), the ping-pong auto-zeroed paths comprising input stages  920  and  925 , feedback stages  930  and  935 , and auto-zero circuits  940  and  945  are dominant, and the amplifier  900  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling currents injected by the auto-zero circuits  940  and  945  ensure that the input-referred offsets of input stages  920  and  925  and feedback stages  930  and  935  do not contribute to the output voltage. 
   At high frequencies, the feedforward path comprising input stage  970  and feedback stage  980  is dominant. Above a threshold frequency, the feedforward path ensures that a feedback signal V fb  can track the input signal V in . As a result, even if mixing occurs due to the gating at the inputs of input stages  920  and  925  and feedback stages  930  and  935 , the resulting mixing products cancel. 
     FIG. 10  illustrates a schematic of a current feedback instrumentation amplifier  1000 , including parallel input stages and a high-frequency feedforward path, in accordance with various embodiments of the present invention. In amplifier  1000 , tranconductance amplifiers  1011  and  1016  together serve as an output stage, such as output stage  910  in amplifier  900 , tranconductance amplifiers  1012 ,  1022 , and  1017  serve as first, second, and third input stages, such as input stages  920 ,  925 , and  970  of amplifier  900 , tranconductance amplifiers  1013 ,  1023 , and  1018  serve as first, second, and third feedback stages, such as feedback stages  930 ,  935 , and  980  of amplifier  900 , and resistors  1061  and  1062  together serve as feedback network, such as feedback network  950  of amplifier  900 . Additionally, tranconductance amplifier  1016 , along with capacitors  1056 - 1057 , serves as a Miller-compensated intermediate stage to amplifier  1000 . Capacitors  1051  and  1052  serve as frequency compensators for the tranconductance amplifier  1011 , thus forming a nested-Miller-compensated output stage. Resistors  1061  and  1062 , together with the reference voltage V ref , generate the feedback voltage V fb  based on the output voltage V out . V fb  is fed back as an input to the feedback tranconductance amplifiers  1013 ,  1023 , and  1018 . 
   Switches  1031 - 1036  and  1041 - 1046 , tranconductance amplifiers  1014  and  1015 , and capacitors  1053  and  1054  function together as a first auto-zero circuit, such as auto-zero circuit  940  of amplifier  900 . Similarly, switches  1071 - 1076  and  1081 - 1086 , tranconductance amplifiers  1024  and  1025 , and capacitors  1058  and  1059  function together as a second auto-zero circuit, such as auto-zero circuit  945 . It should be appreciated that switches  1031 - 1036 ,  1041 - 1046 ,  1071 - 1076 , and  1081 - 1086  may be any of a number of devices capable of performing a switching function. In one embodiment, the switches  1031 - 1036 ,  1041 - 1046 ,  1071 - 1076 , and  1081 - 1086  serve to switch the amplifier  1000  between first and second configurations corresponding to first and second phases of operation. For example, the first configuration may correspond to switches  1031 - 1036  and  1071 - 1076  being closed and switches  1041 - 1046  and  1081 - 1086  being open. Conversely, a second configuration may correspond to switches  1041 - 1046  and  1081 - 1086  being closed and switches  1031 - 1036  and  1071 - 1076  being open. 
   During the first phase, the first input stage and the first feedback stage are auto-zeroed while the second input stage and the second feedback stage perform the amplification functions of amplifier  1000 . Conversely, during the second phase, the second input stage and the second feedback stage are auto-zeroed while the first input stage and the first feedback stage perform the amplification functions of amplifier  1000 . 
   Thus, during the first phase, the inputs of the tranconductance amplifiers  1012  and  1013  are shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb , respectively. Any input offsets of amplifiers  1012  and  1013  cause an offset current that flows into the integrator formed by tranconductance amplifier  1014  and capacitors  1053  and  1054 . The output of this integrator then drives the tranconductance amplifier  1015  to generate a nulling current, which effectively nulls the offset current. 
   Concurrently, V in  and V fb  are applied to amplifiers  1022  and  1023 , respectively, and the summed output current of amplifiers  1022 ,  1023 , and  1025  is coupled with the intermediate stage (i.e., amplifier  1016 ). The amplifier  1000  then operates similar to a traditional current feedback instrumentation amplifier, except that a nulling current injected by amplifier  1025  (which is calibrated in the second phase, discussed below) ensures that the input-referred offset voltages of amplifiers  1022 - 1023  do not contribute to the output voltage. 
   At the end of the first phase, switches  1031 - 1036  and  1071 - 1076  open. As a result, the voltage at the output of the integrator around amplifier  1014  is held so that amplifier  1015  continues nulling the offset current at the outputs of amplifiers  1012  and  1013 . 
   Subsequently, in the second phase, switches  1041 - 1046  and  1081 - 1086  are closed. V in  and V fb  are applied to amplifiers  1012  and  1013 , respectively, and the summed current of amplifiers  1012 ,  1013 , and  1015  is coupled with the intermediate stage (i.e., amplifier  1016 ). The amplifier  1000  then operates similar to a traditional current feedback instrumentation amplifier, except that the nulling current injected by amplifier  1015  insures that the input-referred offset voltages of amplifiers  1012 - 1013  do not contribute to the output voltage. 
   During the second phase, while the first input stage and the first feedback stage are performing amplification functions, the second input stage and the second feedback stage are auto-zeroed. In other words, the inputs of the tranconductance amplifiers  1022  and  1023  are shorted to the input common mode voltage V cmin  and the feedback common mode voltage V cmfb , respectively. Any input offsets of amplifiers  1022  and  1023  cause a corresponding offset current that flows into the integrator formed by tranconductance amplifier  1024  and capacitors  1058  and  1059 . The output of this integrator then drives the tranconductance amplifier  1025  to generate a nulling current, which effectively nulls the offset current. 
   During operation, the auto-zeroing circuits of amplifier  1000  periodically switch amplifier  1000  between the first configuration and the second configuration, ensuring that the input stages and feedback stages are periodically recalibrated. Thus, this “ping-pong” operation ensures that there is continuously an offset-free stage in the signal path. 
   At high frequencies, the feedforward path comprising amplifiers  1017 - 1018  is dominant. Together with the output amplifier  1011 , it forms a regular Miller-compensated two-stage amplifier with approximately 20 dB/dec roll-off. In one embodiment, the frequency at which the feedforward path starts to dominate is:
 
ω pz   =g   1018   /C   1051 ,  (2)
 
(assuming C 1051 =C 1052  and g 1017 =g 1018 ). In a preferred embodiment, this frequency is chosen to be below the clock frequency. Above ω pz , the feedforward path ensures that the feedback signal V fb  can track the input signal V in . As a result, switching transients associated with switching between first and second configurations are suppressed. The lower ω pz , the higher the relative gain of the feedforward path at the clock frequency and its harmonics, and therefore the better the attenuation of such switching transients.
 
   In some cases, a residual offset may appear in amplifiers  400 ,  500 ,  600 ,  700 ,  800 ,  900 , and  1000 . This residual offset may be produced by a number of factors. For example, with reference to  FIG. 4 , due to the finite output impedance of amplifiers  412 ,  413 , and  415 , the offset of the output amplifier  411  may result in an offset current that is not compensated for by the auto-zeroing loop formed by amplifiers  414 - 415 , and therefore causes an input-referred offset voltage. Second, the finite gain in the auto-zeroing loop may result in a residual input-referred offset voltage. Thirdly, due to charge injection at the end of the auto-zeroing phase, the voltage stored on integrator capacitors  453 - 454  may change slightly, resulting in a small error in the nulling current injected by amplifier  415  during the amplification phase. It should be appreciated that similar effects may occur in amplifiers  500 ,  600 ,  700 ,  800 ,  900 , and  1000 . 
   In one embodiment, both residual offset due to the offset of the output stage and the residual offset due to finite gain in the auto-zeroing circuitry can be reduced by adding a current buffer stage  1110 , as shown in  FIG. 11 . Although amplifier  1100  as illustrated in  FIG. 11  does not include a feedforward path or ping-pong circuitry, it should be appreciated that the addition of a current buffer stage in a similar manner may be achieved for amplifiers  500 ,  600 ,  700 ,  800 ,  900 , and  1000 . 
   The current buffer stage  1110  increases the impedance at the input of amplifier  411 . Therefore, the gain in the auto-zero loop is increased and the voltage offset of amplifier  411  results in a smaller offset current. In the embodiment depicted in  FIG. 11 , offset introduced by the current buffer stage  1110  itself is removed by the auto-zeroing process. The current buffer may be implemented as a simple cascode stage. Gain boosting may be applied to further reduce the residual offset. In another embodiment, an actual gain stage may be applied instead of, or in combination with, the current buffer stage  1110 . It should be appreciated that in such a case, an extra dominant pole will be introduced that will require additional frequency compensation. 
   In one embodiment, the residual offset due to charge injection may be kept small by using fully-differential circuitry. Charge injection would then be reduced to charge-injection mismatch. In one embodiment, the offset may be further reduced by using small switches and large integrator capacitors. In yet another embodiment, the transconductance (i.e., g 415 , g 815 , etc.) of the nulling amplifiers  415 ,  615 ,  815 ,  825 ,  1015 , and  1025  may be made smaller than the transconductance of their respective input and feedback amplifiers, so that the voltages at the outputs of the integrators will be larger than the offset voltages at the inputs of the respective input and feedback amplifiers. The smaller the transconductance of the nulling amplifiers, the smaller the input-referred offsets due to given errors in the voltages at the outputs of the integrators. 
   It should be appreciated that the tranconductance amplifiers  412 ,  612 ,  812 ,  822 ,  1012 , and  1022  of  FIGS. 4 ,  6 ,  8 , and  10  have associated amounts of input capacitance. As such, the amplifiers  412 ,  612 ,  812 ,  822 ,  1012 , and  1022  may act as switched-capacitor loads to the signal source V in  because they are periodically discharged during auto-zeroing phases and need to be recharged during amplification phases. In various embodiments, this effect may be reduced by using a pre-charging technique.  FIG. 12  illustrates an input stage  1200  of an amplifier (such as amplifier  400 ) that includes pre-charging circuitry, in accordance with various embodiments of the present invention. It should be appreciated that similar configurations may be used in amplifiers  300 ,  500 ,  600 ,  700 ,  800 ,  900 , and  1000  as well. In  FIG. 6 , the input capacitance of amplifier  412  is depicted by capacitor  1255 . Input stage  1200  includes additional switches  1271  and  1272 , which allow for a “pre-charging” configuration of the input stage  1200 , in addition to the amplification configuration and the auto-zeroing configuration. During the pre-charging phase, switches  431 - 432  and  441 - 442  are opened and switches  1271 - 1272  are closed. As a result, the inputs of amplifier  412  are coupled with a buffered version of the input signal V in  via buffers  1281  and  1282 , so that the current needed to charge the input capacitance  1255  is provided by the buffer amplifiers  1281 - 1282 , rather than by the signal source. In one embodiment, the buffers are  1281 - 1282  are unity gain buffers. Thus, in a subsequent amplification phase, the signal source only needs to provide current to correct for any small offset errors of the buffer amplifiers  1281 - 1282 , rather than the full input voltage V in . It should be appreciated that while an input stage  1200  is depicted in  FIG. 12 , the input capacitances of other tranconductance amplifiers may be pre-charged in a similar fashion. For example, feedback amplifiers  413 ,  613 ,  813 ,  823 ,  1013 , and  1023  may be pre-charged to V fb  to reduce loading from their respective feedback networks. 
   Exemplary Operations in Accordance with Various Embodiments 
   The following discussion sets forth in detail the operation of present technology for reducing effects of offsets in current feedback instrumentation amplifiers. With reference to  FIGS. 13-17 , flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A each illustrate example operations used by various embodiments of the present technology for reducing effects of offsets in current feedback instrumentation amplifiers. Flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A include processes that, in various embodiments, are carried out by circuitry in an integrated circuit. Although specific operations are disclosed in flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A, such operations are examples. That is, embodiments are well suited to performing various other operations or variations of the operations recited in flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A. It is appreciated that the operations in flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A may be performed in an order different than presented, and that not all of the operations in flowcharts  1300 ,  1350 A,  1410 A,  1600 , and  1625 A may be performed. 
     FIG. 13  illustrates a flowchart  1300  of a process for reducing effects of offsets in a current feedback instrumentation amplifier, in accordance with various embodiments of the present invention. At block  1310 , an input stage of the instrumentation amplifier may optionally be pre-charged with a buffered version of the input voltage prior to actually applying the input voltage to the input stage. At block  1320 , a feedback stage is similarly pre-charged with a buffered version of a feedback voltage. It is appreciated that the precharge voltages may vary somewhat from the input and feedback voltages themselves. However, pre-charging in this manner reduces loading of the input and feedback voltages by any input capacitances in the input stage and feedback stage respectively. 
   Block  1330  involves generating an intermediate current based on the input voltage. Block  1340  involves generating a feedback current based on an output voltage of the instrumentation amplifier. It is appreciated that in a conventional instrumentation amplifier, the intermediate current and feedback current would have error components due to input offsets of the input stage and the feedback stage. Thus, at block  1350 , a nulling current is generated based on the offset components. It should be appreciated that generating the nulling current may be achieved in a number of ways. For example,  FIG. 14  illustrates a flowchart  1350 A of a process for generating a nulling current, in accordance with various embodiments of the present invention. At block  1410 , the instrumentation amplifier is switched from an amplification configuration to an auto-zero configuration. It should be appreciated that this may also be achieved a number of ways. For example,  FIG. 15  illustrates a flowchart  1410 A of a process for switching and instrumentation amplifier from an amplification configuration to an auto-zero configuration, in accordance with various embodiments of the present invention. At block  1510 , the output stage of the instrumentation amplifier is decoupled from the input stage in the feedback stage. During this period while the output stage is separated from the other stages of the amplifier, additional circuitry may be employed in order to effectively hold the output of the instrumentation amplifier. At block  1520 , the input stage and the feedback stage are coupled with an auto-zero loop. This auto-zero loop may be substantially as described and shown above, but is not limited as such. Block  1530  then involves a common mode input voltage to the input stage. Similarly, block  1540  involves applying a common mode feedback voltage to the feedback stage. 
   With reference again to  FIG. 14 , block  1420  involves measuring the offset components. In one embodiment, this is achieved using an integrator, but is not limited as such. The nulling current is then generated based on the measured offset components (block  1430 ). At block  1440 , the instrumentation amplifier is switched from the auto-zero configuration back to the amplification configuration. Thus, the instrumentation amplifier continues to compensate for the offsets during the amplification configuration by continuing to inject the nulling current. 
   With reference again to  FIG. 13 , block  1360  involves optionally buffering the intermediate current, the feedback current, the nulling current, or any combination thereof. In one embodiment, this may be achieved through the use of a cascode stage. At block  1370 , a high-frequency path, which may be operated concurrently with the auto-zeroed low-frequency path, is utilized to generate the output voltages at frequencies above a particular frequency (e.g., the clock frequency). In other words, at high frequencies, the high-frequency path dominates the low-frequency auto-zeroed path, and the low-frequency auto-zeroed path dominates the high-frequency path at frequencies below the threshold frequency. 
   Referring now to  FIGS. 16A-16B , flowchart  1600  illustrates another process for reducing the effects of offsets in instrumentation amplifier, in accordance with various embodiments of the present invention. At block  1605 , a first input stage of the instrumentation amplifier is optionally pre-charged with a buffered version of the input voltage. Although not illustrated in flowchart  1600 , a first feedback stage of the amplifier may similarly be pre-charged with a feedback voltage. Again, it is appreciated that the pre-charge voltages may vary somewhat from the input and feedback voltages themselves. However, pre-charging in this manner reduces loading of the input and feedback voltages by any input capacitances in the first input stage and first feedback stage respectively. At block  1610 , a first amplification path is provided via a first sub-circuit in a first configuration of the instrumentation amplifier. Block  1615  then involves compensating for input offsets of the first sub-circuit with a first nulling current. At block  1620  the first nulling current may optionally be buffered. This may be achieved, for example, with a cascode stage. It is appreciated that other currents may be buffered in a similar manner. At block  1625 , a second nulling current of a second sub-circuit of the instrumentation amplifier is calibrated. It should be appreciated that the calibration may be achieved in a number of ways. For example,  FIG. 17  illustrates a flowchart  1625 A for a process of calibrating a nulling current, in accordance with various embodiments of the present invention. Block  1710  involves applying a common mode input voltage to the input stage (i.e., the input stage of the second sub-circuit). Block  1720  involves applying a common mode feedback voltage to a feedback stage (i.e., the feedback stage of the second sub-circuit). The common mode input voltage and the common mode feedback voltage will cause the input stage and the feedback stage to generate currents that correspond to any offsets of the input stage and the feedback stage. Thus, block  1730  involves measuring the offset components of the second sub-circuit. 
   With reference again to  FIGS. 16A-16B , block  1630  involves switching the instrumentation amplifier from the first configuration to a second configuration. In one embodiment, the switching involves switching a first sub-circuit of the instrumentation amplifier from an amplification configuration to an auto-zero configuration and switching a second sub-circuit of the instrumentation amplifier from an auto-zero configuration to amplification configuration. At block  1635 , an input stage of the second sub-circuit is optionally pre-charged with a buffered version of the input voltage. Similarly, a feedback stage of the second sub-circuit may be pre-charged with a buffered version of a feedback voltage. 
   At block  1640 , a second amplification path is provided via the second sub-circuit while the instrumentation amplifier is in the second configuration. At block  1645 , input offsets of the second sub-circuit are compensated for using the second nulling current that was calibrated in block  1625 . At block  1650 , the second nulling current is optionally buffered, for example, using a cascode stage. Block  1655  involves calibrating the first nulling current in the first sub-circuit while the instrumentation amplifier is in the second configuration. In one embodiment, the first nulling current may be calibrated as described above with reference to  FIG. 17 , but is not limited as such. 
   At block  1660 , the instrumentation amplifier is switched from the second configuration back to the first configuration. It should be appreciated that this process of switching between the amplification path provided by the first sub-circuit and the second sub-circuit may be repeated numerous times during the operation of instrumentation amplifier. Such continued switching allows for periodic recalibration of the nulling currents, which ensures that the output of the instrumentation amplifier is free of offset errors. Moreover, this ping-pong operation also ensures that the instrumentation amplifier continually has a path from input to output. 
   At block  1665 , a high-frequency path, which may operate concurrently with the auto-zeroed low-frequency path, may be utilized to generate the output voltage. This path may be used, for example, at frequencies above a threshold frequency. In one embodiment, the high-frequency path is separate from the first sub-circuit and second sub-circuit of the instrumentation amplifier. 
   Thus, embodiments provide technology allowing for instrumentation amplifiers with very low input-referred offset, low input current, and low level spurious switching signals at the output. Moreover, some embodiments use a ping-pong architecture, which ensures that there is constantly an offset-free stage in the signal path, and no additional offset is thereby introduced due to aliasing. Additionally, spurious signals may be further reduced by adding a high-frequency feedforward path. 
   The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.