Patent Publication Number: US-8537063-B2

Title: Antenna for reception of satellite radio signals emitted circularly, in a direction of rotation of the polarization

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a U.S. application and hereby claims priority from German application Serial No. 10 2009 011 542.0 filed on Mar. 3, 2009 the disclosure of which is hereby incorporated herein by reference in its entirety. 
     One embodiment of the invention relates to an antenna for reception of satellite radio signals emitted circularly, in a direction of rotation of the polarization. 
     With satellite radio signals, efficiency is important, both with regard to the transmission output emitted by the satellite and with regard to the efficiency of the reception antenna. Satellite radio signals are generally transmitted with circularly polarized electromagnetic waves, because of polarization rotations on the transmission path. In many cases, program contents, for example, are transmitted in frequency bands that lie very close to one another, in terms of frequency, as shown in  FIG. 1 . Using the example of SDARS satellite radio, this happens at a frequency of about 2.33 GHz, in two adjacent frequency bands each having a bandwidth of 4 MHz with a distance between the center frequencies of 8 MHz. The signals are emitted by different satellites, with an electromagnetic wave that is circularly polarized in one direction. Accordingly, for reception, antennas circularly polarized in the corresponding direction of rotation are used. Such antennas are known, for example, from DE-A-4008505 and DE-A-10163793. This satellite radio system is additionally supported by the transmission of terrestrial signals, in certain regions, in another frequency band having the same bandwidth, disposed between the two satellite signals. Similar satellite radio systems are currently in the planning stage. 
     The antenna known from DE-A-4008505 is built up on a conductive base surface oriented essentially horizontally, and consists of crossed horizontal dipoles having dipole halves inclined downward in V shape and consisting of linear conductor parts, which halves are fixed in place mechanically, relative to one another, at an azimuthal angle of 90 degrees, and are attached at the upper end of a linear vertical conductor attached to the conductive base surface. The antenna known from DE-A-10163793 as well as U.S. Pat. No. 6,653,982 wherein this antenna is also built up above a conductive base surface that is generally oriented horizontally, and consists of crossed frame structures mounted azimuthally at 90 degrees relative to one another. In the case of both antennas, the antenna parts that are spatially offset by 90 degrees relative to one another, in order to produce the circular polarization, are interconnected offset by 90 degrees relative to one another in the electrical phase. 
     For example  FIG. 1A  discloses a flat antenna for mobile satellite communications with circular polarization. Here, two antennas whose planes  0  are orthogonal to one another are combined in a particularly advantageous embodiment, wherein each antenna, has an asymmetrizing network  9  and a matching circuit  17 . At the output of matching circuit  17 , the voltage U z  for circular polarization is formed by means of a phase-rotation element  18 , and a summation circuit  19 . The latter, as shown in  FIG. 1A , are constructed by connecting in parallel, lines whose lengths differ by λ/4. This antenna also includes an antenna collection point  11 , an impedence or capacitor  7 , a vertical symmetry axis  8 , and a conductor portion  16 . 
     Both forms of antennas are particularly suited for reception of satellite signals, which are emitted by high-flying satellites—so-called HEOS. However, the signals of geostationary satellites—of so-called GEOS—come in at a lower elevation angle in the regions at a distance from the equatorial zones. The reception of such signals is possible only with a comparatively small antenna gain, in the case of the two antenna forms mentioned, and therefore is problematical, because of the weak transmitter power of the satellites—which results from economic considerations. In addition, there is the difficulty of structuring the antennas to have a small construction height, which is necessarily required, particularly for mobile applications. Patch antennas are known as other antennas of this type, according to the state of the art, but these are also less efficient with regard to reception at a low elevation angle. 
     One embodiment of the invention is configured to indicate an antenna having a low construction height, which is particularly suitable also for efficient reception of satellite signals emitted in circularly polarized manner in a direction of rotation, which signals come in at low elevation angles. 
     This task is accomplished, in the case of an antenna according to the preamble of the main claim, by means of the characterizing characteristics of the main claim and the measures proposed in the other claims. 
     Furthermore, an antenna of this type can advantageously be combined, in a common construction space, with antenna structures that also receive a circularly polarized field, and can be used, together with these antenna structures, in an antenna diversity system or a system for digital beam shaping with azimuthal beam scanning. This combination is particularly interesting also for reception systems in which signals from GEO satellites and HEO satellites are supposed to be received in the same manner in closely adjacent frequency bands. In this connection, the antenna combination is characterized by a particularly low reciprocal coupling of the antennas with one another. 
     SUMMARY 
     According to one embodiment, the antenna for reception of circularly polarized satellite radio signals comprises at least two antenna elements connected with an antenna connector, which antenna elements are linearly polarized in a spatial direction, in each instance, and are connected by way of an matching and phase shifter network wherein one of the antenna elements is formed as a loop antenna, essentially disposed in a horizontal plane. The loop antenna for its electrically effective shortening, has at least one interruption bridged by a capacitor. The loop antenna can have multiple interruptions disposed at a distance from one another and bridged by capacitors. At least one interruption of the conductor loop forms a loop antenna connection point of the loop antenna. Furthermore, at least one additional antenna element is present, which demonstrates linear polarization and is connected with its antenna element connection point as well as with the loop antenna connection point, by way of an matching and phase shifter network, which network is configured so that with reciprocal operation of the antenna as a transmission antenna, the radiation fields of the loop antenna and of the at least one additional antenna element are superimposed with different phases in the far field of the antenna. This at least one additional antenna element can have a polarization oriented perpendicular to the polarization of the loop antenna. All the antenna elements can be essentially formed from thin, wire-shaped conductors of similar conductor structures. 
     For the production of antennas that are known from DE-A-4008505 and DE-A-10163793 (also published as U.S. Pat. No. 6,653,982 to Lindenmeier) the individual antenna parts are placed on planes that intersect at a right angle, and that these planes additionally stand perpendicular on the conductive base plane. This particularly holds true for the frequencies of several gigahertz that are usual for satellite antennas, for which particularly great mechanical precision is necessary in mass production of the antennas, in the interests of polarization purity, impedance matching, and reproducibility of the directional diagram. The production tolerances that are required for antennas according to one embodiment of the present invention can be adhered to significantly more easily, in advantageous manner. Another very significant advantage of at least one embodiment results from the property that in addition to the horizontally polarized loop antenna, at least one additional antenna element is present, which has a polarization oriented perpendicular to the polarization of the loop antenna. This antenna element can advantageously be used, when terrestrial, vertically polarized transmission signals are present, also for reception of these signals. 
     The distribution of the currents on an antenna in reception operation is dependent on the terminating resistance at the antenna connection point. In contrast to this, in transmission operation, the distribution of the currents over the antenna conductors, with reference to the feed current at the antenna connection location, is independent of the source resistance of the feeding signal source, and is therefore clearly linked with the directional diagram and the polarization of the antenna. Because of this clear situation in combination with the law of reciprocity, according to which the radiation properties—such as directional diagram and polarization—are identical in transmission operation and in reception operation, the task according to at least one embodiment of the present invention, with regard to polarization and radiation diagrams, is accomplished on the basis of the configuration of the antenna structure to produce corresponding currents in transmission operation of the antenna. In this way, the task according to at least one embodiment of the present invention is also accomplished for reception operation. All the considerations below, concerning currents on the antenna structure and their phase reference point, therefore relate to reciprocal operation of the reception antenna as a transmission antenna, unless reception operation is specifically addressed. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       In the following, the invention will be explained in greater detail using exemplary embodiments. The related figures show, in detail: 
         FIG. 1A  shows a prior art FIG. relating to a flat antenna for mobile satellite communications; 
         FIG. 1B  shows a graph of frequency bands of two satellite radio signals with circularly polarized radiation, in the same direction of rotation, in close frequency proximity; 
         FIG. 2  discloses a schematic perspective view of a first embodiment of an antenna; 
         FIG. 3  discloses an antenna as in  FIG. 2 , but with a plurality of monopoles disposed with rotation symmetry relative to a center axis Z; 
         FIG. 4  discloses an antenna as in  FIG. 2 , but with a loop antenna with two antenna connection points formed to lie opposite one another; 
         FIG. 5  discloses an antenna as in  FIG. 4 , whereby, however, conductor parts of the loop antenna have been used to form the rotation-symmetrical roof capacitor; 
         FIG. 6  discloses an antenna similar to  FIG. 2 , but with a vertical feed line to supply the loop antenna, whereby the feed line additionally forms a vertical monopole and the loop antenna forms a roof capacitor of the monopole; 
         FIG. 7  discloses an antenna similar in design to that shown in  FIG. 6 , but with a loop antenna formed as a square, with the center Z; 
         FIG. 8  discloses another embodiment of the invention with phase-differential superimposition of the reception voltages from the horizontal and the vertical electrical field components of a loop antenna and a monopole antenna formed by the vertical two-wire line; 
         FIG. 9  discloses a design similar to that shown in  FIG. 2 , whereby in place of discrete capacitors, the capacitor, which is formed from a circuit of multiple reactive elements, in each instance, in such a manner that at different frequencies, different capacitance values are in effect; 
         FIG. 10  discloses a combined antenna array of another embodiment for separate availability of LHCP signals and RHCP signals, respectively, of different satellite signals at different antenna connection points with a vertically polarized monopole configured as a rod antenna, a horizontally polarized loop antenna and a 90 degrees hybrid coupler; 
         FIG. 11  discloses an antenna array similar to that shown in  FIG. 10 , but with implementation of the monopole according to the antenna array in  FIG. 6 , by means of a combination of the effects of the loop antenna as a roof capacitor and the two-wire line; 
         FIG. 12  discloses another embodiment showing an alternative uncoupling of RHCP signals and LHCP signals, respectively, for diversity technologies, controlled by a changeover switch situated in a radio reception module; 
         FIG. 13  discloses another embodiment showing diversity technologies, with LHCP/RHCP changeover switch as in  FIG. 12 , but, similar to the antenna in  FIG. 8 , without a separate monopole; 
         FIG. 14  discloses a design similar to that shown in  FIG. 5 , but with a common antenna element connection point for the common feed of the loop antenna and of the vertical monopole with roof capacitor; 
         FIG. 15A  discloses the Vertical directional characteristics of the LHCP-polarized electromagnetic field of an antenna as shown in  FIG. 2 , with circular polarization at low elevation angles and with azimuthal independence of the phase of the radiation; 
         FIG. 15B  discloses the vertical direction characteristics of the LHCP polarized electromagnetic field of an antenna of  FIG. 2  with a crossed antenna element according to the state of the art, or, respectively, of a ring line antenna element according to the invention as in  FIG. 19 , with circular polarization at high elevation angles; whereby the phase of the circular polarization turns with the azimuthal angle of the propagation vector; 
         FIG. 16A  discloses a vertical directional characteristic of the LHCP-polarized electromagnetic field of an antenna for 2.3 GHz, corresponding to  FIG. 18 ; 
         FIG. 16B  discloses horizontal directional characteristic of the LHCP-polarized electromagnetic field at an elevation angle of about 30 degrees with minimal radiation for the azimuthal angle of 180 degrees; 
         FIG. 17  discloses another embodiment comprising a loop antennae with two symmetrically disposed loop antenna connection points and monopole with construction space marked in the center Z for a crossed antenna element; 
         FIG. 18  discloses a schematic diagram of a perspective view of an antenna as shown in  FIG. 17 , having a centrally affixed crossed antenna element with a new type of ring line antenna element for producing a circularly polarized field with an azimuthally dependent phase; 
         FIG. 19  discloses a perspective view of a ring line antenna element; 
         FIG. 20  discloses an antenna as in  FIG. 18 , but for production of the continuous line wave with a λ/4 there is a coupling conductor conducted at an advantageous distance—with reference to the line wave resistance—parallel to the ring line antenna element; 
         FIG. 21  discloses an antenna as in  FIG. 20 , but with a λ/4 directional coupler; 
         FIG. 22  discloses a perspective view of a loop antenna configured in square shape, and a ring line antenna element formed as a closed, square line ring having an edge length of λ/4; 
         FIG. 23  discloses a perspective view of another embodiment with a square ring line antenna element  7   c  as in  FIG. 22 ; 
         FIG. 24  discloses a perspective view of another embodiment; 
         FIG. 25  discloses a perspective view of a circular group antenna element; and 
         FIG. 26  shows a perspective view of another embodiment; 
     
    
    
     DETAILED DESCRIPTION 
     One embodiment is directed to a reception antenna, wherein the following properties of the antenna will be described for reciprocal operation of the antenna as a transmission antenna, for reasons of better reproducibility, whereby the transmission case, however, applies also for the directional diagrams of the reception case, because of the reciprocity relationship that of course applies. 
     In the following, the fundamentals concerning configuration of antennas, on which the antenna according to the invention is based, are explained. 
     Turning in detail to the drawings,  FIG. 1B  shows for example a graph of frequency bands of two satellite radio signals with circularly polarized radiation, in the same direction of rotation, in close frequency proximity. These two bands are LHCP bands designated with the first band having a center frequency at fmu and the second band having a center frequency at fmo. 
     The particular advantage of an antenna according to the embodiment as shown, for example, in  FIG. 2 , is the property that the electrical field intensity vector produced in the far field during operation of the antenna as a transmission antenna, in accordance with the reciprocity law, also describes a substantially pure, in the technical sense, circular polarization with an azimuthal all-around characteristic even at relatively low elevation angles of the radiation.  FIG. 2  shows a schematic perspective view of a first embodiment of an antenna a loop antenna  14  over the conductive base surface  6 , with horizontal polarization and with a monopole  7   a  configured as a rod antenna as an additional antenna element  7  in the center axis Z of the horizontal loop antenna  14 . The additional antenna element  7  for reception of vertically polarized fields. The antenna has a matching network  25  and phase shifter network  23  for phase-differential superimposition of the reception of the horizontally and vertically polarized field components in a summation network  53 . 
     With a phase-rigid combination of the horizontally polarized loop antenna  14  with the at least one vertical antenna element  7 , there is superimposition of the remote radiation fields of the two antenna elements by 90 degrees, by means of correspondingly different phase feed and corresponding amplitude feed of the two antennas. With this, two field intensity vectors are produced in the remote radiation field, in a plane perpendicular to the propagation direction, which vectors stand perpendicular on one another and differ by 90 degrees in phase, and represent the desired circularly polarized field. For the production of the all-around characteristic, it is required that the phase reference points B—also called phase emphasis points—of the two antennas coincide, and this is achieved by means of a rotation-symmetrical placement about the common center axis Z of the antennas. 
     Antenna  14  can be a circular or polygonal loop antenna  14  disposed above the base surface  6  in a plane having a constant distance  4  as a height h. This antenna acts essentially similar to a frame antenna over a conductive surface. With the prerequisite of an azimuthally constant current application to the loop antenna  14 , the elevation angle of the main radiation direction can be adjusted by way of the selection of the height h and the horizontal expanse—this means the radius in the case of a circular configuration of the loop antenna  14 . In this connection, a zero position in the vertical direction and in the horizontal direction can be achieved. Achieving a desired vertical directional diagram, however, requires a horizontal expanse of the loop antenna in such a manner that its total circumferential length is no longer small in comparison with the electrical free space wavelength λ 0 . According to this embodiment of the invention, the loop antenna is therefore divided into n equal line sections having the length Δs&lt;λ 0 /8, by means of interruption points  5 , which are connected with one another, in each instance, by means of insertion of a capacitor such as capacitor  16 . In this connection, the capacitors are preferably selected so that together with the properties of the line sections, resonance occurs at the operating frequency fm. Such an antenna can advantageously be configured for an azimuthally substantially pure omnidirectional characteristic. In combination with the at least one vertical antenna element  7 , which is present in the center Z of the loop antenna  14  in the example of  FIG. 2 , and whose azimuthal radiation diagram is also omnidirectional, the desired circularly polarized radiation field with a substantially pure all-around characteristic is also obtained for the antenna of  FIG. 2 . Thus, this type of antenna is advantageously suitable particularly for satellite radio reception in vehicles, where antennas having an azimuthal all-around characteristic are applied to the electrically conductive external skin of the vehicle. 
       FIG. 2  shows a circular loop antenna  14  having a radius R, which antenna can also be configured to be polygonal. Its phase reference point B is located at its center point, at the center axis Z. The structure is subdivided into “n” line sections, each having the length Δs. The total circumference length is S. The antenna acts as a frame antenna having dimensions in the range of the wavelength, whereby nevertheless, according to the invention, a homogeneous current distribution is achieved by means of subdivision of the structure and insertion of capacitors  16 . As a result, the antenna acts electrically shortened in length, and produces a homogeneous, horizontally polarized electromagnetic field all around. The loop antenna  14  is disposed at a constant height h above the conductive base surface  6 . The vertical main radiation direction can be set by way of the selection of the height h and the radius of the loop antenna  14 . A zero position can be achieved in the vertical direction and in the horizontal direction. 
     The ring-shaped circumferential line length S is divided into n equally long pieces having a length Δs=S/n. Let the line wave resistance of the circumferential line above the conductive base surface  6  be Zw. The capacitative reactance ΔX per line piece Δs and thus the capacitance value C=1*ΔX) to be inserted into this line piece, in each instance, is defined, assuming an extended length Δs and with an approximately ring-shaped line having a large radius R of the ring-shaped loop antenna  14  relative to the line height h, by
 
Δ X/Zw =tan(2πΔ s/λ   0 ).
 
     The capacitance value C to be inserted into the line piece Δs is obtained in a good approximation:
 
 C= 1/(ω Zw  tan(2πΔ s/λ   0 ))
 
     Circular frequency of the satellite signals=ω; free space wavelength of the satellite signals=λ 0    
     With this dimensioning of the capacitance values C, resonance can be set for the loop antenna  14 , so that the antenna impedance that occurs at the loop antenna connection point  3  can be configured to be real, to a great extent. 
     To obtain an all-around diagram, in a good approximation, the line having the length S must be divided into sufficiently many partial pieces by inserting capacitors  16 . For a reasonable division, the following applies: Δs/λ 0 &lt;⅛. If the partial pieces Δs=S/n are selected to be sufficiently small such as Δs/λ 0 &lt;⅛, then equality Δs of all the partial pieces is not absolutely necessary, as long as a capacitor  16  is inserted after every partial piece, the value of which capacitor is calculated according to the criterion described above, from the relative length Δs/λ 0  of the partial piece in question. 
     As an additional antenna element  7 , in the example of  FIG. 2 , an electrically short, vertically oriented monopole  7   a  is affixed in the center Z of the loop antenna  14 . The deviation of the positioning of the monopole  7   a  from the center Z should not exceed λ 0 /20, in the interests of roundness of the radiation diagram. At an interruption point of the loop antenna  14 , its loop antenna connection point  3  is formed, at which, by way of a two-wire line  26 , an matching network  25  with balance-unbalance elements  29  and a subsequent phase shifter network  23  are connected. The antenna element connection point  2  of the monopole  7   a  follows the matching network  25  for impedance adjustment, and the signals of the monopole  7   a  and of the loop antenna are superimposed in the summation network  53 ; this, in turn, is connected with the antenna connection point  28 . To produce the circularly polarized radiation, the phase of the phase shifter network  23  and all the networks, in their interaction, are set in such a manner that the radiation fields of the loop antenna  14  and those of the monopole  7   a  are superimposed, in the far field of the antenna, at a phase difference of 90 degrees and with the same intensity. 
     To avoid non-symmetries of the azimuthal directional diagram of the monopole  7   a , brought about by the two-wire line  26 , which runs essentially vertically, the latter is structured, according to the invention, so that it acts with an inductive high ohm effect, with regard to the longitudinal current that flows in common mode, which is superimposed on the current pair that flows on the two conductors in push-pull mode. In this way, the result is achieved that the two-wire line  26  does not influence the radiation field of the monopole  7   a . A number of possibilities exists for the configuration of such a two-wire line  26 . In practice, it can be produced in advantageous manner, for example, by means of a two-wire line printed onto a carrier, which line is configured as a meander, in order to increase its inductance. In addition, a desired phase relationship can be produced by means of the selection of its length. 
     By way of different weighting in the superimposition of the two antenna signals, the vertical directional diagram can be filled out in the direction of low elevation angles for these signals. The monopole  7   a , configured as a rod antenna, possesses a similar main radiation direction, in its vertical directional characteristic, as the horizontally polarized loop antenna  14 , but makes a bigger contribution than the latter for low elevation angles. Using the networks  25 ,  23 ,  53 , not only can the weighting of the properties of the two antenna signals be set differently, but also the required phase condition can be met. 
     The influence of a symmetrical vertical feed line in the form of the symmetrical two-wire line  26  not situated in the center axis Z does not reduce the polarization purity of the loop antenna  14  itself. It is advantageous if the connection of the one connector on the non-symmetrical side of the matching and balance-unbalance element  25 ,  29  with the further circuit of the antenna array takes place using a microstrip conductor  30  passed over the conductive surface  6 . The other connector on the non-symmetrical side of the balance-unbalance element  29  is connected with the electrically conductive base surface  6 . Because of the symmetry properties of the two-wire line  26 , the effects of the currents that flow in opposite directions on the conductors of the two-wire line  26  compensate one another to a sufficient degree, so that these also do not influence the radiation properties of the loop antenna  14 . As will be explained in the following, the currents produced on these conductors by the electromagnetic reception field also have no influence on the effects at the antenna connection point  3 . With regard to the azimuthal radiation diagram of the monopole  7   a , however, residual non-symmetry can occur, as a function of the radius R of the loop antennas  14 . 
     It corresponds to the nature of the present invention that both the axis relationship and the spatial orientation of the ellipsis for elliptical polarization can be set by means of setting the matching networks  25  and the phase shifter network  23 . This settability can be utilized in very advantageous manner, according to the invention, for example, in antenna diversity technologies, to continuously optimize the reception power in the reception field distorted by multi-path propagation, by means of current matching of the ellipticity of the polarization. 
     As an example for the configuration of reception in the range of an elevation angle between 25 degrees and 65 degrees (typical angle range for GEO-stationary satellite reception) with an azimuthal all-around characteristic, a horizontally disposed loop antenna  14  is placed above the conductive base surface  6  at a distance of about 1/10 of the wavelength. It is advantageous if the diameter of the loop antenna  14  is selected to be not significantly smaller than ¼ of the wavelength. Along the conductor path, at intervals of about ⅛ of the wavelength, an interruption point  5  equipped with a capacitor  16  having a reactive resistance of about −200 Ohm is inserted. Because of the effect of the capacitors  16  according to the invention, it is possible to achieve an azimuthally constant current distribution on the loop antenna  14 , as required for all-around radiation, although the extended length of the loop antenna  14  is not short in comparison with the wavelength λ. On the other hand, this length is necessary in order to bring about a practicable impedance of the loop antenna  14 . In  FIG. 15(   a ), the vertical diagram of such an antenna according to this embodiment is shown as an example. For the example of a square-shaped loop antenna  14  with a central short vertical monopole, in the frequency range around 2.3 GHz, an edge length of about 3 cm and a height h of 13 mm have proven to be advantageous for the loop antenna  14 , to implement both the vertical directional diagram according to  FIG. 15(   a ) and a matching conductor wave resistance Zw. 
     Another property that should be emphasized as compared with the antennas known from the state of the art, such as, for example, those from DE-A-4008505 and DE-A-10163793 and which issued as U.S. Pat. No. 6,653,982 titled Flat Antenna for Mobile Satellite Communication, which issued on Nov. 25, 2003 and wherein the disclosure of which is hereby incorporated herein by reference. In contrast to this, the phase of the antennas mentioned above, according to the state of the art, changes with the azimuthal angle of the propagation vector, in other words by the angle 2π in the case of a complete azimuthal revolution. The significance of these properties, according to one embodiment of the invention, with regard to a combination of antennas according to the cited state of the art with an antenna according to one embodiment of the invention will be explained below. 
     For the case that the satellite radio system is additionally supported by means of transmission, in certain regions, of vertically polarized terrestrial signals, in another frequency band having a similar bandwidth, closely adjacent in frequency, it is desirable to fill up the vertical directional diagram for the vertical component of the electrical field intensity toward low elevation angles. The connection, according to the invention, of the loop antenna  14  and the additional antenna element  7 , polarized vertical to it—implemented, in most cases, as a vertical monopole—makes it possible to take this aspect into consideration in particularly advantageous manner. 
     In  FIG. 3  an antenna according to one embodiment of the invention is shown, whereby the additional antenna element  7 , which is oriented perpendicular on the plane of the loop antenna  14 , is formed from a group of monopoles  7   a . These are disposed with rotation symmetry relative to the center Z and within the loop antenna  14 . The monopoles are connected with one another at their lower end, by way of lines, at the center Z, and form the antenna element connection point  2  there. If the diameter of the circular ring on which the monopoles  7   a  are disposed around the center Z is not too great, and if the number of monopoles  7   a  is not too low, the azimuthal directional diagram of the antenna element  7  configured in this manner is sufficiently omnidirectional. 
       FIG. 4  shows another embodiment of an antenna similar to  FIG. 2 , whereby the loop antenna  14  has two antenna connection points  3   a ,  3   b  that lie opposite one another in the plane of symmetry SE, in order to reduce the residual non-symmetry of the array with regard to the azimuthal directional diagram of the monopole  7 , at which points the balance-unbalance- and matching networks  25 ,  29  disposed in the loop plane are connected. These outputs are switched in parallel by way of equal phase shifter networks  23  and connected with the two-wire line  26 . The additional antenna element  7  disposed in the center axis Z is structured as a monopole  7   b  with horizontal conductor parts, disposed with rotation symmetry relative to the center axis Z, as a roof capacitor. These conductor parts are also structured to be symmetrical to the plane of symmetry SE. 
       FIG. 5  shows another embodiment similar to  FIG. 4 , wherein the conductor parts of the loop antenna  14 , however, is used to form the rotation-symmetrical roof capacitor  12 . In the case of a completely symmetrical configuration of the roof capacitor  12  both with regard to rotation symmetry and also similar to the plane of symmetry SE shown in  FIG. 4 , the function of the loop antenna  14  is not impaired by the connector of the roof capacitor  12  of the monopole. 
     In  FIG. 14 , an antenna according to the invention is shown as in  FIG. 5 , but with a common antenna element connection point  2  for common feed of the loop antenna  14  and of the vertical monopole with roof capacitor  7   b . The circularly polarized field occurs in that the waves that arrive at the loop antenna  14  in transmission operation, by way of the vertical monopole antenna and by way of the horizontal arms of the roof capacitor  12 , split up to right and left, whereby the distance to the nearest capacitor  16  on the loop antenna toward the right side is selected to be different from the distance to the nearest capacitor  16  on the loop antenna toward the left side. The loop antenna must therefore be rotated about the z axis, relative to the roof capacitor, in such a manner that different angle distances α and β occur on the left side and the right side, between the horizontal arms of the roof capacitor and the next capacitor, in each instance. In this manner, a loop antenna connection point for feeding the ring current to the loop antenna  14  is formed, in interaction of the feeding horizontal arms of the roof capacitor  12  and the related interruptions of the conductor loop. In this connection, the horizontal arms of the roof capacitor  12  are fed by way of the antenna element connection point  2 , not only for its effect as a roof capacitor, but furthermore also for production of the ring current on the loop antenna  14 , so that feed of the loop antenna  14  and of the monopole  7   b  with roof capacitor can take place in efficient, very advantageous manner, by way of the common antenna element connection point  2  of the monopole  7   b.    
       FIG. 6  shows another advantageous embodiment of the invention according to the functional principle of the antenna in  FIG. 2 , but with a vertical feed line  26  disposed in the center Z, for feeding the loop antenna  14 , whereby the feed line  26  forms a vertical monopole  7   a  and the loop antenna  14  forms a roof capacitor  12  of the monopole  7 . The loop antenna  14  is formed with two antenna connection points  3   a ,  3   b  disposed symmetrically relative to one another, and an matching network  25  in the loop plane, in each instance, as well as with a central connection to the vertical feed line to the matching network  33 , configured as a two-wire line  26 . In this connection, the effects of the currents of the loop antenna  14  that flow in opposite directions on the conductors of the two-wire line  26 , in push-pull mode, compensate one another. The reception voltage of the monopole  7   a  is passed to the power splitter and phase shifter network  31  at its antenna element connection point  2 , as a common mode of the two-wire line  26 , at one output, and the reception voltage of the loop antenna  14  is passed to this network as a push-pull mode of the two-wire line  26 , at the other output of the matching network  33 , for amplitude-appropriate and phase-differential superimposition of the signals at the antenna connector  28 . 
       FIG. 7  shows another advantageous embodiment of the antenna according to the functional principle of the antenna in  FIG. 6 , but with a loop antenna  14  configured as a square with the center Z, which antenna is formed by four dipoles  21  disposed in a square, which lie horizontally and are connected at their ends by way of capacitors  16 , with a distribution network  10  disposed centrally in the phase reference point B and connected by way of feed lines  18 . The dipole system acts as a roof capacitor of the vertical monopole formed in this manner, similar to that explained in  FIG. 5 . The reception of horizontal and/or vertical electrical field components takes place by way of sum formation  34  or difference formation  35 , respectively, and phase-differential superimposition of the signals by way of the phase shifter network  23  and the summation network  53 . 
     In another advantageous embodiment of the invention, in  FIG. 8 , an antenna array is shown with phase-differential superimposition of the reception voltages from the horizontal and the vertical electrical field components of a loop antenna  14  and a monopole antenna  7   a  formed by the vertical two-wire line  26 . Similar to  FIG. 4 , here again, two antenna connection points  3   a ,  3   b  with matching networks  25  are present in the plane of the loop antenna  14 , which points lie opposite one another in the plane of symmetry SE, to improve the symmetry of the array. Using a dipole network  61  introduced into one of the conductors of the two-wire line  26 , setting of the common mode to push-pull mode ratio takes place on the vertical two-wire line  26 , thereby setting the ratio of the proportion of the vertically polarized field at low elevation of the main radiation direction to the proportion of the horizontally polarized field at higher elevation of the main radiation direction. In addition, the setting of the phases required for production of circular polarization takes place using this summation network  53 . According to the invention, the axis relationship and the spatial orientation of the ellipsis for elliptical polarization can be set by means of the selection of the aforementioned common mode to push-pull mode ratio and the phase setting. 
     In another advantageous embodiment of the invention in  FIG. 9 , the antenna—for example similar to the embodiment as in FIG.  2 —is configured, however, as a multi-frequency range antenna. For this purpose, in place of discrete capacitors in the loop antenna  14 , the capacitors  16  are formed, in each instance, from equal dipole networks, consisting of a circuit of multiple reactive elements, in each instance. In this way, different capacitance values are in effect at different operating frequencies, which values allow the resonance for configuring the real antenna impedance at these different operating frequencies. 
     In  FIG. 1B , the situation is shown that two satellite radio frequency bands having a small bandwidth Bu and Bo (typically about 4-25 MHz), respectively, are emitted closely adjacent, at a high frequency in the L band and in the S band, respectively, in any case at a frequency of fm&gt;1 GHz, with the same directions, in other words with left-rotating circular polarization (LHCP), for example. At a bandwidth Bu and Bo, respectively, of a few megahertz (typically about 4-25 MHz), the relative frequency interval between the center frequencies fmu and fmo is so slight that frequency-selective configuration of the antenna is not possible, and not necessary if the frequency bandwidth of the antenna is suitable. Both signals can therefore be received at the same antenna connection point  28 , because of the sameness of the directions of rotation of the polarization. For the case that another satellite radio signal is present in the close frequency vicinity, having the other circular polarization, this can be configured by means of configuring two separate antenna connection points  28   a  and  28   b  within the scope of a combined antenna according to the invention.  FIG. 10  shows an antenna array having a vertically polarized monopole  7  configured as a rod antenna, and a horizontally polarized loop antenna  14  according to the invention, with a common phase reference point B with regard to the transmission case, but with separate feed of the signals to the connector for vertical polarization  49  and to the connector for horizontal polarization  48 , respectively. The hybrid coupler  45  with 90 degrees positive and negative phase difference with regard to the LCHP connector  28   a  and the RCHP connector  28   b , respectively, which coupler follows these connectors, allows separate availability of LHCP signals and RHCP signals, respectively, having different directions of rotation of the circular polarization. The monopole  7 , structured as a rod antenna  32 , has an interruption point  5  connected with a reactive element  8 , to configure its vertical diagram. 
     For the reception of geostationary satellites, in particular, whose signals come in at a comparatively low elevation in northern latitudes, it is provided that the one essentially perpendicular monopole  7  contains at least one interruption point  5 , which is connected, i.e. bridged with at least one reactive element  8 , to configure the vertical diagram. In this manner, the vertical diagram can advantageously be adapted to the requirements. The antenna connection point  2  is formed at the foot point of the monopole  7 , at the connector to the matching network  33 . 
     A similar antenna array is shown in  FIG. 11 , whereby the implementation of the monopole  7 , however, takes place similar to the antenna array in  FIG. 10 , by means of the combination of the loop antenna  14  that acts as a roof capacitor, and the two-wire line  26 . Using a combined matching circuit  50 , not only the matching of the loop antenna  14  and the matching of the monopole  7 , but also the setting of a common phase reference point B are brought about. 
     In another advantageous antenna array for alternative uncoupling of RHCP signals and LHCP signals, respectively, as shown in  FIG. 12 , a loop antenna  14 —as in FIG.  6 —is provided with two antenna connection points  3   a ,  3   b  and matching networks  25  connected with them and situated in the loop plane, which networks are implemented, for example, as λ/4 transformation lines. The outputs of the matching networks  25  are switched in parallel to add up. The reception signal is passed to an matching network  25  situated on the base surface  6 , by way of the two-wire line  26 , the output of which network is connected, in turn, to one of the two inputs of a signal combination circuit, particularly configured as a 90 degrees hybrid coupler  45 . At the antenna connection point  2  at the foot point of the monopole  7   a , situated in the center Z of the array and configured as a rod antenna, a matching network  25  is also connected, the output of which network feeds the other of the two inputs of the 90 degree hybrid coupler  45 . An LHCP/RHCP changeover switch  55  connected with the outputs of the 90 degree hybrid coupler  45  makes satellite reception signals of the two rotation directions of the polarization available alternatively, at the connection point  28 —controlled by a changeover control situated in a radio receiver module  52 . In the case of control with a diversity control module  38  situated in an LHCP/RHCP radio module  52 , the antenna array can advantageously be used also for polarization diversity, by means of switching between reception for LHCP waves and RHCP waves. 
     In another particularly efficient embodiment of such an antenna having a circularly polarized field with a reversible direction of rotation, in FIG.  13 —similar to the antenna in FIG.  12 —the separate monopole  7  is eliminated. The reception in the case of vertical polarization is brought about by the two-wire line  26 . The phase difference that is required for superimposition of the reception signals of the loop antenna and of the monopole is brought about by the network  61 . By means of inserting a suitably configured dipole network  61  into one of the strands of the vertical two-wire line  26 , the difference of 90 degrees between the phases of the horizontal field component absorbed by the vertical two-wire line  26  with the loop antenna  14  as a roof capacitor  12  and by the loop antenna  14  is set in such a manner that their combination is present at the microstrip conductor  30  to the matching network  54  with this phase difference, and thus also at the connection point  28 . As a result, the antenna receives a circularly polarized field. A circuit that links the reception signals of the loop antenna  14  at the output of the matching networks  25  from the horizontally polarized electrical field and the reception signals of the vertical two-wire line  26  from the vertically polarized electrical field comprises an LHCP/RHCP changeover switch  55  for changing the polarity of the reception voltage of the loop antenna  14 . In this manner, the latter voltage can be added to the reception voltage from the vertically polarized electrical field, with a different sign, so that it is possible to switch between reception of the LHCP field and of the RHCP field by means of switching the LHCP/RHCP changeover switches  55 . Controlled by a changeover control between LHCP and RHCP reception signals situated in the receiver, signals having differently rotated polarization of the satellite signals, from different transmission paths, are alternately available. 
     As has already been explained in connection with the antenna in FIG.  8 —here again, a corresponding network  61  composed of reactive resistors can be interconnected with the strand of the vertical two-wire line  26  that is connected with ground. Using the network  61 , the setting of the common mode to push-pull mode ratio on the vertical two-wire line  26  can be set. The reception voltages from the horizontal and the vertical electrical field components are superimposed in phase-differential manner, in accordance with the circular polarization. By means of setting the common mode to push-pull mode ratio on the vertical two-wire line  26 , the ratio of the proportion of the vertically polarized field at a lower elevation of the main radiation direction can be set relative to the proportion of the horizontally polarized field at a higher elevation of the main radiation direction. 
     In another particularly advantageous embodiment of the invention, the antenna in the above embodiments is combined with another antenna element having an azimuthal all-around diagram, whose polarization is circular, and the phase of the circular polarization rotates with the azimuthal angle of the propagation vector—in other words with a complete azimuthal rotation about the angle 2π. As already mentioned above, the antennas known from DE-A-4008505, DE-A-10163793, and EP 1 239 543 B1, respectively, from the state of the art, as well as other known antenna forms, fulfill this requirement. The method of effect of these antennas is essentially based on that the individual antenna parts are placed on planes that cross at a right angle and stand perpendicular to the base plane, and that the antenna parts of the different planes are interconnected offset in phase by 90 degrees, to produce the circular polarization. Even the effect of patch antennas can be represented in similar manner. Antenna elements  7   d  having an azimuthal all-around diagram, whose polarization is circular and whose phase of circular polarization rotates with the azimuthal angle of the propagation vector, and which are composed of two crossed antenna elements, will be referred to as “crossed antenna elements” in the following, for a simpler differentiation. 
     In the case of a combination of such a crossed antenna element  7   d , in such a manner that its phase reference point B coincides with the antenna according to the invention as described above, and the signals of the two antennas are combined in amplitude-appropriate manner, by way of a controllable phase rotation element  39  and a summation network, a main direction of the radiation occurs, in advantageous manner, in the azimuthal directional diagram of the combined antenna array, which direction is dependent on the setting of the phase rotation element  39 . 
     The method of effect of the superimposition of the signals will be explained using  FIGS. 15 and 16 . In  FIG. 15   a , the vertical directional characteristic of the LHCP-polarized electromagnetic field of an antenna according to the invention, as described above, is shown. The phase of this field is independent of the azimuthal angle and thus the phase for the azimuthal angles 0 degrees and 180 degrees, in each instance, is characterized with the same angle—0 degrees in the example. In comparison with this, the elevation directional diagram of another antenna element  7   d  described above is shown in  FIG. 15   b , of a type as produced by a crossed antenna element  7   d  as described above, whereby different phase values result for the azimuthal angles 0 degrees and 180 degrees, differing by 180 degrees, which are characterized with 0 degrees and 180 degrees in the example. Thus, in the case of phase-equal superimposition of the two signals, the antenna gain of the combined antenna array can be increased for the azimuthal angle 0 degrees and weakened for the azimuthal angle 180 degrees, and a zero point of the directional diagram can actually be set, in the case of a suitable setting of the amplitudes at a desired elevation angle, as shown in  FIG. 16 . If the two signals are superimposed offset relative to one another by the adjustable phase angle φ, then—based on the phase change of the circular polarization of the crossed antenna element ( 7   d ) with the azimuthal angle of the propagation vector—the azimuthal directional diagram is obtained, while maintaining the elevation directional diagram about the same angle φ, rotated in the one direction or the other. In this manner, the directional diagram of the combined antenna array can advantageously be tracked with its main direction pointing to the satellite, for example, in mobile operation, or an interference, for example, can be blocked out, in targeted manner, by means of assigning a direction to the zero point of the directional diagram. Particularly in the case of satellite reception in vehicles, in this way the signal/noise ratio can always be structured optimally, within the scope of dynamically tracked setting of the directional diagram. 
     In  FIG. 17 , the combined antenna array according to the invention is shown with a crossed antenna element  7   b  indicated by the construction space  42 , as it is shown, for example, in EP 1 239 543 B1, there in  FIG. 10   a . In this connection, the vertical antenna conductor  20  indicated there is structured in equivalent manner here in  FIG. 17 , as a vertical monopole  7   a  in the center Z, and is uncoupled from the connection point  56  of the crossed antenna element  49  because of the symmetry conditions. The latter is connected with the summation network  53  by way of the controllable phase rotation element  39 , in which network the signals of the loop antenna  14 , of the vertical monopole  7   a , and of the crossed antenna element  49  are combined, with the suitable weighting, in each instance, to form the reception signal of the combined antenna array. In equivalent manner, an antenna of the type as described in DE-A-4008505, or a patch antenna having the vertical monopole  7   a  in the center Z, as well as an array of dipoles crossed parallel above the ground surface can be combined. 
     All arrays of n equal horizontal antenna element elements  59  can be used for this, if they are disposed in such a manner that their centers produce the corners of an equilateral polygon, and if rotation of the array about the z axis, by an angle of 360 degrees/n, reproduces the structure in itself, and if the feed of antenna element elements that are adjacent in the direction of rotation, in each instance, differs in phase by 360 degrees/n, in each instance. In  FIG. 25 , such arrays are shown for the example of four and five antenna element elements, in each instance. with n equal horizontally polarized antenna element elements  59  disposed with rotation symmetry about the center Z, whose feed differs in phase by 360 degrees/n, in each instance, from antenna element elements adjacent in the direction of rotation.  FIG. 25  top: n=4.  FIG. 25  bottom: n=5. 
     In a particularly advantageous further development of the invention, in place of an antenna element of the “crossed antenna element” type as described, a new type of additional antenna element  7   c  according to the invention, with circular polarization and an azimuthal all-around radiation diagram, the phase of which rotates with the azimuthal angle of the propagation vector, referred to in the following as a ring line antenna element  7   c , for differentiation, is used. In  FIG. 15(   b ), the vertical diagram of such an antenna according to the invention is shown as an example. 
     According to one embodiment of the invention, the ring line antenna element  7   c  is configured as a polygonal or circular closed ring line disposed with rotation symmetry about the center Z, running in a horizontal plane at the height h 1  above the conductive base surface  6 . 
     According to one embodiment of the invention, the ring line is fed in such a manner that the current distribution of a running line wave occurs on it, the phase difference of which wave over a rotation amounts to precisely 2π; thus the extended length of the ring line corresponds to the wavelength λ that occurs on the ring line. The radiation contributions of the horizontally polarized individual line sections are superimposed in the far field, in such a manner that the desired radiation with circular polarization and the required phase dependence on the azimuthal propagation direction and the essentially omnidirectional azimuthal directional characteristic occur. In the case of a circular configuration of the ring line, its horizontal expanse is therefore D=λ/π. In the case of a ring line as shown in  FIG. 18 , the wavelength λ on the ring line is equal to the free space wavelength λ 0 . To reduce the diameter D, the wavelength λ on the ring line can take place by increasing the line inductance or/and the line capacitance relative to the conductive base surface  6 . This can take place in known manner, for example preferably by means of introduction of concentrated inductive elements into the line structure, or, for example, by means of a meander-shaped structure of the ring conductor. 
       FIG. 18  discloses a schematic diagram of a perspective view of an antenna as shown in  FIG. 17 , having a centrally affixed crossed antenna element with a new type of ring line antenna element for producing a circularly polarized field with an azimuthally dependent phase of feed at ring line feed points  20   a ,  20   b  spaced apart from one another by λ/4, of signals that differ in phase by 90, to produce a rotating wave having a wavelength over the circumference of the line. 
     This combined antenna array, consisting of the loop antenna  14  and the monopole  7   a  combined with it at a phase difference, to produce the circularly polarized radiation field with azimuthally independent phasing and a circular ring line antenna element  7   c  disposed concentrically, with the center Z, having a ring line connection point  19  for superimposition of its circularly polarized radiation field, but with azimuthally dependent phasing and for control of the azimuthal main direction by way of the controllable phase rotation element  39 . The phase emphasis of the ring line antenna element  7   c  lies in the center Z of the antenna array, as a consequence of the phase distribution on the rotation-symmetrical ring line structure described, and thus coincides with the phase reference point B of the loop antenna  14  and that of the monopole  7   a , as described—independent of the position of the controllable phase rotation element  39 . Production of the continuous line wave on the ring line antenna element  7   c  takes place, proceeding from the ring line connection point  19 , by way of the power splitter and phase shifter network  31 , at whose outputs signals offset in phase from one another by 90 degrees are applied, which signals are connected, by way of an matching network  25 , in each instance, by way of the feed lines  18 , to ring line feed points  22   a  and  22   b  along the ring line structure, which are spaced apart from one another by λ/4. The particular advantage is connected with a ring line antenna element  7   c  of this type that it is configured concentric to the loop antenna  14  and with a greater diameter in comparison with it. A crosswise dimension usual for the loop antenna  14  can be configured within broad limits, but is generally smaller than λ/4 and can therefore be structured within the ring line antenna element  7   c  having the diameter λ/π. This allows advantageously generous configurability of the vertical monopole  7   b  or monopole system, respectively, situated in the center Z, as in  FIGS. 3 ,  4 , and  5 , for example. Because of the geometrically required radiation uncoupling between the loop antenna  14  and the ring line antenna element  7   c  that surrounds it, the diameters of the two antenna elements can be configured, within broad limits, independent of one another, in the interests of the configuration of their vertical directional diagrams and the resulting vertical directional diagram of the antenna array at the antenna connector  28 . Likewise, the distance h of the plane of the loop antenna  14  from the conductive base surface  6  can be selected to be different from the distance h 1  between the plane of the ring line antenna element  7   c  and the conductive base surface  6 , although it is particularly efficient for production if both antenna elements are imprinted onto the same planar carrier, for example in printed form. In  FIG. 16(   a ), the vertical diagram, and in  FIG. 16(   b ), the horizontal diagram of such an antenna according to the invention are shown as examples. For the example of a loop antenna  14  having a square shape, with a central, short vertical monopole, in combination with a ring line antenna element also having a square shape, in the frequency range about 2.3 GHz, an edge length of about 3 cm and a height h of 13 mm for the loop antenna  14 , and an edge length of about 3.4 cm, which corresponds to about ¼ of the wavelength, and a height h of 10 mm for the square-shaped ring line antenna element have proven to be advantageous for implementing not only the directional diagram according to  FIGS. 16A and 16B . 
     The loop antenna  14  is connected with the summation network  53  for forming the circularly polarized radiation, with azimuthal independence of the phase, by way of the two-wire line  26  that is at high ohms for common mode currents, by way of an matching network  25 , and the monopole  7   a  is connected with it by way of an matching network  25  and by way of the phase shifter network  23 . Likewise, the ring line connection point  19  is connected with the summation network  53  by way of the controllable phase rotation element  39 , and the signals are superimposed on the other signals there, with the suitable weighting to produce the desired vertical directional diagram of the antenna array with an adjustable azimuthal main direction at the antenna connection  28 . 
       FIG. 19  discloses a perspective view of a ring line antenna element  7   c , but fed by way of four feed points  22  offset by λ/4, in each instance, along the ring line, by signals offset in phase by 90°, in each instance. The feed sources can be obtained in known manner, by means of power splitting and 90° hybrid couplers  45 . 
     This design is to complete the azimuthal symmetry wherein the feed sources can be obtained in known manner, by means of power splitting and 90 degree hybrid couplers  45 . 
     In an advantageous embodiment of the invention, production of the continuous line wave on the ring line antenna element  7   c  takes place analogous to  FIG. 18 , but by means of the λ/4 coupling conductor  43  in  FIG. 20 . The latter is guided at an advantageous distance, with regard to the line wave resistance, over an extended length of λ/4, parallel to the ring line antenna element  7   c . For production, the λ/4 coupling conductor  43  can economically be applied to the same carrier as the ring line antenna element  7   c  and, if applicable, the loop antenna  14 , in printed form. 
     In another advantageous embodiment of the invention, production of the continuous line wave on the ring line antenna element  7   c  takes place analogous to  FIG. 20 , but by means of λ/4 directional coupler  44  in  FIG. 21 . A λ/4 coupling conductor  43  is guided parallel to a microstrip conductor  30 , and, together with the λ/4 coupling conductor  43  coupled with the ring line antenna element  7   c , forms the λ/4 directional coupler  44 . 
     In  FIG. 22 , the ring line antenna element  7   c  of an antenna is configured similar to  FIG. 18 , but as a closed, square line ring above the conductive base surface  6 , having the edge length of λ/4 in a plane at the distance h 1  above the conductive base surface  6 . Coupling to the ring line antenna element  7   c  takes place in contact-free manner, by way of the λ/4 coupling conductor  57  configured in ramp shape, with the ring line connection point  19 . 
     Likewise, the loop antenna  14  with its capacitors  6  is disposed, as a square conductor structure, within the ring line antenna element  7   c , with the same center Z. The other antennas are not shown, for reasons of clarity. In  FIG. 22 , the ramp-shaped λ/4 coupling conductor  43  should be emphasized as a particularly advantageous form of contact-free coupling to the ring line antenna element  7   c.    
     Proceeding from the ring line connection point  19  situated on the conductive base surface  6 , a vertical feed line  18  leads all the way to one of the corners, except for a coupling distance  58 , in order to meet with the base surface  6  there, essentially in accordance with a ramp function, below an adjacent corner, and to be connected with this surface in electrically conductive manner. This form of coupling is particularly advantageous for efficient production, since because of the square structure of the ring line antenna element  7   c , the ramp-shaped  214  coupling conductor  43  can be configured on a planar carrier. By means of setting a suitable coupling distance  58 , impedance matching at the ring line connection point  19  can furthermore be brought about in advantageous manner. 
     In  FIG. 23 , the ring line antenna element  7   c  is also configured to be square, as in  FIG. 22 , but is fed at its corners, in each instance, by way of a feed line  18 , which runs over an equal length, as a microstrip conductor  30 , on the conductive base surface  6 , in each instance, and contains a vertical conductor of equal length, in each instance. The other antennas are not shown, for reasons of clarity. The feed line  18 —proceeding from the ring line connection point  19 —are connected with a power distribution network that consists of microstrip conductors  30  ( 15   a ,  15   b ,  15   c ) switched in a chain and having a length of λ/4. The wave resistances of the microstrip conductors  30 —proceeding from a low wave resistance at the ring line connection point  19 —at which one of the feed lines  18  is directly connected—are stepped up in such a manner that the signals fed into the ring line antenna element  7   c  at the corners possess the same power values and differ in phase by 90 degrees, in each instance, continuously trailing one another. The other antenna parts are also not shown, for reasons of clarity. 
     In another embodiment as shown in  FIG. 24 , another antenna element in the form of an outer ring line antenna element  7   e  is present. This embodiment discloses a loop antenna  14 , monopole  7   a , ring line antenna element  7   c  and the additional outer ring line antenna element  7   d , on which a continuous line wave of two wavelengths is produced, to raise the emission gain by increasing the emission bundling. 
     In contrast to the ring line antenna element  7   c , the circumference of which precisely corresponds to a wavelength λ—in other words a full period—the circumference of the outer ring line antenna element  7   e  is selected to be two wavelengths λ, so that in the case of excitation by signals displaced in phase by 90 degrees relative to one another, at ring line feed points  22  along the outer ring line structure, spaced apart from one another by λ/4, a continuous line wave occurs on the ring line antenna element  7   d . This feed occurs, in the example in  FIG. 24 , in the case of both ring lines, in similar manner, by way of the matching networks  25  and the power splitter and phase shifter network  31 . The connection point  21  of the outer ring line antenna element  7   e  is also connected with the summation network  53 , so that the effects of the radiation of the outer ring line antenna element  7   e  occur as a function of the weighting at the antenna connection  28 . The signals at the loop antenna/monopole connection point  27 , at the ring line connection point  19 , and at the connection point  21  of the outer ring line antenna element  7   e , are combined in weighted manner in the summation network  53 , by way of controllable phase rotation elements  39 , so that an increased antenna gain is achieved at the antenna connection  28 , in the azimuthal main direction that is set. Because of the greater diameter of the outer ring line antenna element  7   e , its contribution has a more sharply bundling effect than that of the circularly polarized ring line  7   c . Although the polarization is no longer purely circular by adding the outer ring line antenna element  7   e , the emission gain for certain situations can be increased by means of this measure, because of the overall sharper bundling. 
       FIG. 26  shows a perspective view of a circular group antenna element  7   f  according to an array as in  FIG. 25 , with horizontally polarized antenna element elements  59  disposed at the corner points of a square having the center Z, with feed lines  18  and power splitter and phase shifter network composed of microstrip conductors  30  having a length of λ/4, with the partial pieces  15   a ,  15   b ,  15   c.    
     With this design, a circular group antenna element  7   f  of the type described in  FIG. 25  is shown. This consists of multiple horizontally polarized antenna element elements  59  disposed in a plane parallel to the conductive base surface  6  and at a distance from it, and azimuthally about the center Z, with rotation symmetry, on a circle K. By way of feed lines  18  with phase shifter network, a common circular group antenna element connection point  60  is created. In the case of reciprocal operation of the antenna, excitation of the circular group antenna element  7   f  is brought about in such a manner that each antenna element element  59  is excited with a current having the same amplitude, but in terms of phase, in such a manner that the amount of the current phase is selected to be equal to the azimuth angle Φ of the azimuthal position of the antenna element element  59 , proceeding from an azimuthal reference line, so that the current phase rises or falls with an increasing azimuth angle Φ. For this purpose, the horizontally polarized antenna element elements  59  are disposed at the corner points of a square having the center Z, and are oriented perpendicular to the connection lines between the corner point in question and the center Z, in each instance. The horizontally polarized antenna element elements  59  are connected with the connectors of a power splitter and phase shifter network by way of a feed line  18  of equal length, in each instance. This network is configured from microstrip conductors  30  having a length of λ/4, interconnected in a chain on the conductive base surface  6 , and having the partial pieces  15   a ,  15   b ,  15   c , whose wave resistances—proceeding from a low wave resistance at the circular group antenna element connection point  60 —to which one of the feed lines  18  is directly connected—are stepped up in such a manner that the signals fed into the antenna element elements  59  at the corners possess the same power values and differ in phase by 90°, in each instance, continuously trailing one another. 
     Accordingly, while a few embodiments of the present invention have been shown and described, it is to be understood that many changes and modifications may be made thereunto without departing from the spirit and scope of the invention as defined in the appended claims.