Patent Publication Number: US-11646765-B2

Title: Wireless resonance coupled energy transmission

Description:
FIELD OF THE INVENTION 
     The current invention relates to wireless energy transmission by means of inductive or capacitive coupled resonant circuits. 
     Further, the current invention relates to the control of the resonance frequency, to control the power to be transmitted, and to electrically control the coupling of coupled resonant circuits. 
     BACKGROUND OF THE INVENTION 
     Wireless power supply devices can be realized by means of inductive and/or capacitive proximity coupling. This is used in many RFID systems and wireless battery chargers. In this case, a source unit generates an alternating electromagnetic field. This alternating electromagnetic field is coupled through coupled coils (inductive coupling) or by an open capacitor (capacitive coupling) to a load, in the following referred to as a load unit. With increasing distance from the source unit to the load unit decreases the coupling strength and reduces the receivable amount of power at the load unit. In the case of an open capacitor thereby minimizing the coupling capacity, and in the case of coupled coils thereby increasing the leakage inductance. It is known that this effect can be compensated in case of the leakage inductance by a compensation capacitor and in the case of the coupling capacitor by a compensation inductor. This results in at least one resonant circuit at the source unit side and at least one resonant circuit at the load unit side of the power transmission system. These resonant circuits compensate the leakage inductance and coupling capacitance under the condition that the resonant circuits are tuned exactly to the same resonance frequency and the source unit of the power transmission system operates at this resonance frequency. Such coupled resonant circuits are the base of band filters and have been used for many years for e.g. the coupling of amplifier stages etc. 
     In “Wireless Power Transfer System Description” of the “Wireless Power Consortium (WPC)”, a resonant circuit is shown, which is driven by a generator. Here, a plurality of part inductors is selectively used in a resonant circuit in order to concentrate the radiated energy field to the surface where load units are placed. Further, the power supply is controlled by an upstream voltage or current regulator. 
     The general disadvantage of wireless power principles, based on coupled resonant circuits in the source- and/or load unit, is the resonance frequency detuning due to component tolerances, component aging, coupling and load changes. This detuning effect is undesirable because the impedance of the resonant circuit is frequency selective and a predetermined operation frequency does no longer coincide with the resonant frequency of the circuit. Consequently, the power transmission system operates no longer toward a real load resistance, but also toward an inductive or capacitive component. Thus, the resulting reactive power increases the power dissipation. This lowers the overall efficiency in the source unit (resonant circuit driver stage, etc.) and thus reduces the efficiency of the entire power transmission link. In addition, distortions increase, because the driver circuits generate more harmonics. 
     The known network sensing method measures the resonance frequency of the network during a time interval and operates the system at this resonance frequency thereafter, but the system has no ability to control the resonance frequency of the resonant circuit or network actively. This would be very desirable, due to guidelines such as EN300330, REC7003 and ITU-RSM2123, which determines maximum power levels over frequency ranges (e.g. 119 . . . 135 kHz). 
     Furthermore, there exist national specific constraints for narrow frequency ranges within a frequency band that require much lower power level limits. It is therefore important to control not only the power level but also the spectral position of the emitted power. 
     Another problem relates to the product of the variable coupling (k) and the quality (Q) in the further description referred to as energy coupling (k·Q). The quality factor Q is a measure of the energy stored in the system and the energy transferred by the system, or in other words a measure of the reactive power circulating in the resonant circuit and output power of the resonant circuit. The coupling (k) is essentially determined by the geometry of the wireless coupling link such as distance (area, distance), its angular orientation and the coupling medium. A change in the load resistor in a resonant circuit will also change the energy coupling (k·Q). The system is more or less damped and the energy coupling (k·Q) is consequently smaller or larger. A desired constant output voltage or a desired constant output current of the load unit requires therefore the control of the output power in the source unit by means of a data-load modulation link and/or the control of the output voltage and/or current on the load unit side. 
     “A Frequency Control Method for Regulating Wireless Power to Implantable Devices” proposes frequency detuning in the source unit for the output voltage regulation. The problem with this solution is the detuning of the resonance frequency that also simultaneously controls the power of other coupled load units if more than one load unit is used in the wireless power transmission link. An independent control is not possible. Thanks to the performed detuning on the load side, several load units are independently adjustable. However, disadvantageously, the source unit is no longer loaded with a real load resistance in the wireless power transmission link in both variants. The energy transfer is no longer based on coupled resonant circuits and the overall network does not operate in the real domain (resonance case). Consequently, this causes higher losses in the source unit and/or in the load unit. 
     In “Wireless Power Supply for Implantable Biomedical Device Based on Primary Input Voltage Regulation”, the output voltage of the load unit is digitized and transmitted to the source unit. The operating voltage of the source unit is controlled based on the received data of the load unit. This approach is pursued in the standard of “Wireless Power Consortium (WPC)”, wherein a wireless power transmission link up to 5 watts is specified. This approach is efficient because only the amount of power is transmitted as needed. Unfortunately resonance detuning is not considered. 
     Another fundamental problem is not considered in all known systems, and relates to the upper energy coupling boundary value (k·Q=1), wherein the maximum possible power can be transmitted without any substantial frequency detuning or without substantially bandwidth increase. This case is very important because a small coupling factor (k) can be compensated with a higher quality (Q). In this manner, the energy coupling (k·Q) can be held constant. Thus, for example, the power transmission distance can be increased. 
     In practice, however, often results in a dynamic coupling (time-varying coupling (k)) and/or dynamic load. Examples are different wireless power transmission links with dynamic coupling due to variable geometries, and/or variable distance between the source unit and load unit, and/or modifying the coupling media and especially by altering the load resistance. 
     It is an essential desire to design a wireless power transmission system as universal as possible and ideally approximating a wired connection as much as possible. This would automatically result in the best efficiency and additionally serves maximal flexibility. 
     It is further desirable that for a given source unit the operational distance of the power transmission link can be optimized in designing the load unit without any design modifications in the source unit (coils and/or capacitors). E.g., a load unit featured with a small coupling (k) (small coupling surface, etc.) can compensate its limited range with a greater quality factor (Q). This would meet requirements of an open flexible standard, wherein only the basics shall be specified. 
     Additionally, multiple arbitrary load units coupled under changing coupling conditions (changing coupling factors (k) and/or changing qualities (Q)) shall maintain operable in parallel on a source unit. 
     Further, it should be possible to specify the transmission frequency, as there are radiation limits by law (amplitude and frequency), such as EN300330, REC7003 and ITU-RSM2123. 
     Further it should be possible to vary the transmission frequency in order to reduce the spectral peak power. 
     The following invention describes methods and their implementation details that meet the mentioned requirements. The methods described in the following invention and its detail implementations are featured by low-cost implementations, stable operation and high efficiency. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIGS.  1   a    . . .  1   g  show basic simplified equivalent circuits of a near field wireless power transmission system using coupled resonant circuits and their transmission properties. 
         FIG.  2    shows the block diagram of a wireless power transmission system based on resonance coupling according to a first embodiment of the current invention. 
         FIG.  3    shows the block diagram of an wireless power transmission system based on resonance coupling according to a second embodiment of the current invention. 
         FIG.  4    shows waveforms of output signals of over coupling detectors in accordance with  FIGS.  2  and  3   . 
         FIGS.  5   a    . . .  5   h  show universal embodiments of one coupling inductance used in two main applications according to the current invention. 
         FIGS.  6 A and  6 B  show waveforms and a block diagram of a communication controller according to  FIGS.  2  and  3   . 
         FIG.  7    illustrates a package structure or communication protocol according to  FIGS.  2  and  3   . 
         FIG.  8    shows a detailed circuit according to the current invention. 
         FIGS.  9 A and  9 B  show waveforms according to the  FIG.  8   . 
         FIG.  10    shows a further detailed circuit according to the current invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG.  1   a    shows a wireless power transmission system. A source couples energy via two coupled coils L 1  and L 2  to a load RL. A leakage compensation network is implemented on the source- and load unit side to compensate the leakage inductance. 
       FIG.  1   b    shows the wireless power transmission link according to  FIG.  1   a    with a simplified transformer equivalent circuit  11  and the capacitors C 1  and C 2  as leakage compensation. C 1  and C 2  form together with the primary and secondary inductances L 1  and L 2  the coupled resonant circuits having substantially the same resonance frequencies. C 1  is tuned with L 1  to a primary resonance frequency. C 2  is tuned with L 2  to a secondary resonance frequency. A parallel resonant circuit, formed by C 1  and L 1  on the primary side and a series resonant circuit formed by C 2  and L 2  on the secondary side couple a current source I 1  to a load resistor RL. The primary leakage inductance  12  with a value of L 1 −M, the secondary leakage inductance  13  with the value of L 2 −M and the main inductance  14  with the value M correspond to the coupled coils. The values M, coupling k, primary inductance L 1  and secondary inductance L 2  are related by the equation M=k·√{square root over (L 1 ·L 2 )}. The secondary quality (Qs) is determined by 
               Qs   =         ω   ·   L     ⁢   2     RL       ,         
wherein ω is the angular frequency. As smaller RL is, the greater is Qs.
 
       FIG.  1   c    shows the wireless transmission link according to  FIG.  1   a    with a simplified transformer equivalent circuit  11  and the capacitors C 1  and C 2  as leakage compensation. C 1  and C 2  form together with the primary and secondary inductances L 1  and L 2  the coupled resonant circuits having substantially the same resonant frequencies. C 1  is tuned with L 1  to a primary resonance frequency. C 2  is tuned with L 2  to a secondary resonance frequency. A series resonant circuit formed by C 1  and L 1  on the primary side and a parallel resonant circuit formed by C 2  and L 2  on the secondary side, coupling a voltage source V 1  to a load resistor RL. The value M is determined equal the one shown in  FIG.  1   b   . The secondary quality (Qs) is determined by Qs=RL·ω·C 2 . As greater RL is, the greater is Qs. 
     The transmission characteristics of the equivalent circuits of  FIGS.  1   b  and  1   c    are identical, they differ only in their resonant circuit topologies. These resonant circuit topologies can be combined arbitrarily. E.g., the resonant circuit in the source unit and the resonant circuit in the load unit can have the same topology and can be applied to the described invention as further embodiments. The absolute value of the transfer function I 2  versus I 1  according to  FIG.  1   b    respectively V 2  versus V 1  according to  FIG.  1   c    is shown in  FIG.  1   d   . The phase response of the input impedance respectively the input admittance of the wireless power transmission link is shown in  FIGS.  1   e    . . .  1   g.    
     Curve  1  shows the transfer characteristics for energy coupling (k·Q) smaller than one. The amount of the transfer function reaches a small maximum value and the phase function has a continuous slope during the phase transition from +90 to −90 degrees (see  FIGS.  1   d  and  1   e   ). This characteristic is called undercritical coupled. 
     Curve  2  shows the transfer characteristics for energy coupling (k·Q) approximately equal to one. The amount of the transfer function reaches the maximum value without dip and the phase function has a continuous slope with a minor zero gradient range (without changing the pitch sign) within the phase transition from +90 to −90 degrees (see  FIGS.  1   d  and  1   f   ). This characteristic is called critically coupled and determines the upper energy coupling boundary value. 
     Curve  3  shows the transfer characteristics for energy coupling (k·Q) greater than one. The amount of the transfer function reaches two maximum values and the phase function undergoes a sign change in the slope gradient within the phase transition from +90 to −90 degrees (see  FIGS.  1   d  and  1   g   ). This characteristic is called overcritical coupled and is avoided in the wireless power transmission link in accordance with a first aspect of the invention. In the topology according to  FIG.  1   b   , it is easy to see that the energy coupling (k·Q) increases responsive to a decrease in RL. This corresponds to a load increase which acts toward the power source. 
     Load coupling by means of a series resonant circuit in the load unit is therefore suitable for a voltage source power supply operation (a desired constant output voltage on the load unit side requires a low source impedance). In fact, a greater load (smaller RL) couples itself tighter to the source unit due to the greater energy coupling. Similar is true for  FIG.  1   c   . Here, the energy coupling (k·Q) increases when RL increases. This corresponds to a load increase which acts toward the power source. 
     Load coupling by means of a parallel resonant circuit in the load unit is therefore suitable for a current source power supply operation (a desired constant output current on the load unit side requires a high source impedance). In fact, a greater load (larger RL) couples itself tighter to the source unit due to the greater energy coupling. Regardless of the resonant circuit topology in the source- and load unit the source unit on the primary side sees a transformed load resistance that damps the resonant network. The resulting overall quality (Qtot) of the wireless transmission path determines the efficiency and frequency selection of the source unit. A high Qtot represents a high frequency selection, large reactive power with respect to output power and therefore a rather lower efficiency. A low Qtot represents a low frequency selection, a small reactive power with respect to output power and therefore a rather higher efficiency. 
       FIG.  2    depicts a block diagram of a source unit  200  and a load unit  250  in accordance with a first embodiment of the current invention. An energy source  201  supplies an amplifier  202 .  202  drives a parallel resonant circuit formed by L 1  and C 1 . L 1  is the primary coil of the wireless power transmission link. A zero- or sign detector  203  generates an output signal indicative of the zero crossing of the resonance voltage. 
     A zero- or sign detector  203   a  using a current sensor  203   b  generates a signal indicative of the zero crossing of the resonance current.  203  and  203   a  are substantially identical. The output signal of  203   a  supplies a function controller  206 . An output signal of  203  is coupled to a data demodulator  205 .  205  operates substantially phase- or frequency sensitive and couples demodulated data from the load unit (FSKLdata) to  206 . The output signal from  203  is also coupled to block  206  and to a period processor  204 . 
       204  analyzes input signals time- or frequency sensitive and generates in relation to a predetermined comparison value, an output signal (OCT), which in turn serves  206  as an input signal. An output signal of  206  controls the frequency of a generator  207 , which drives  202 . For that purpose comprises  206  a phase comparator with a subsequent low pass filter. 
     The phase comparator controls the phase difference of the output signals  203  and  203   a  by means of block  207  to zero (resonance frequency). For example, a phase difference at the output of the phase comparator inside  206  increases or decreases the control voltage, which controls  207 . This increases or decreases the frequency of the output signal of  207  until the phase difference reaches again zero (resonance frequency). Another output of  206  controls the output voltage or the output current of block  201 . Consequently, the resonant circuit energy respectively the radiated power of the wireless transmission link is controlled. 
     The secondary coil L 2  of the wireless transmission link forms together with C 2  a series resonant circuit, which is tuned to fsoll (corresponding to the resonant frequency of L 1  and C 1 ). 
     A rectifier  251  rectifies the resonance voltage and couples it via a low pass filter comprising Lf, Cf to a switch SW. When SW is closed, the load resistance RL is coupled through a storage inductance Ls to the charging capacitor Cf. If SW is open and a current flows in Ls, RL is not coupled with Cf and the diode D forming a freewheel circuit. 
     The configuration SW, D, Ls, Cs and RL corresponds to a voltage step-down converter. A step-down controller  255  measures the output voltage across RL and generates at its output essentially a pulse width signal (PWM signal) to control SW. For this purpose  255  comprising at least: a voltage reference, a comparator, a step-down loop filter and a PWM generator (not shown). A zero detector  252  signals a sign change in the resonance voltage. 
     Alternatively, in a further embodiment, the resonance current or the resonance voltage in the resonant circuit L 2 , C 2  is detected (not shown). Exact zero crossing detection is not absolutely necessary, it is essential to obtain accurate phase signals of the oscillating circuit period (or half period). The output signal from  252  serves  255  (the contained PWM generator), an analog/digital converter  256  and a data modulator  253  the synchronization signal. Additionally, the output signal from processor  252  is coupled to a period processor  254 . 
       254 , corresponding to  204 , analyzes the input signal time- or frequency sensitive and generates in relation to a predetermined comparison value, an output signal (OCL) that controls block  255 . One or more output voltages (e.g. the voltage across Cf and/or RL) are in one or more blocks  256  analog to digital converted and then in  253  compared versus a reference value (feedback versus reference comparison). A resulting control difference value is converted in  253  to an appropriate transmission format and along with other data, which are generated and/or stored in  253 , coupled as FSKLdata to the ModSW. ModSW modulates by means of Cm the load of  250 . FSKLdata is transferred in this manner to  200 . Alternatively, as another embodiment, ModSW controls C 2  by means of Cm (not shown). 
     The output voltage control or regulation of the current invention can be carried out in various modes. The modes described below are individually and/or jointly active at different times as a combination in various embodiments.  255  signals the selected mode over the connection mode to block  253 . This mode selection signal controls  253 , so that the relevant control data (control difference value) of the respective operation mode are transmitted. 
     In one mode, SW remains for the output voltage regulation always closed or  250  does not include a voltage step down converter (RL is directly connected to Cf). Thus,  256  detects the output voltage across RL. The control difference value (feedback to reference value comparison, i.e. comparison with a first voltage reference value) in FSKLdata corresponds to the output voltage deviation from a desired output voltage across RL. This control difference value is serially transmitted via the wireless coupling link to block  205 .  206  uses this data in a standard control function (I, PI or PID) and controls block  201 . In this manner the control loop is closed from RL over FSKLdata toward block  201 . 
     In another mode,  250  comprises only SW but no voltage step down converter. The voltage regulation in  250  is implemented by simply interrupting the energy flow (opening SW) in  250  and/or is implemented as described above over the closed loop via the control difference value in FSKLdata and block  206  in  200 . 
     In another mode, SW operates as a PWM controller in the step-down converter for the output voltage regulation in  250 .  256  detects the voltage across Cf. The control difference value (feedback to reference value comparison, i.e. comparison with a second voltage reference value) in FSKLdata corresponds to the voltage deviation from a desired voltage across Cf. This control difference value is transmitted together with a duty cycle information of SW (its PWM signal or a digital signal corresponding to the PWM duty cycle of SW) serially over the wireless coupling link to block  205 . 
       206  uses this data in a standard control function (I, PI or PID) and controls block  201 . In this manner the control loop is closed from RL over FSKLdata toward block  201 . This permits to optimize the efficiency of the wireless power transmission link and yet ensures the operation of several load units  250  on one source unit  200 . Additionally, the fast loop response (shortest loop) in the output voltage regulation via SW results in the best dynamic properties. 
     A larger output voltage control range is obtained when the voltage step-down converter is combined with a voltage step-up converter. This is implemented as a further embodiment of the current invention and is shown in  FIG.  2    with the dashed components. For this purpose, a diode Ds is coupled in series with the switch SW, and a second switch SWs placed in parallel to the diode D. 
     Advantageously, D and SWs are combined. In the case  250  operates as a voltage step-down converter (voltage across Cf is higher than the desired output voltage across RL),  255  act as described above. SW defines by its PWM the output voltage and SWs can be left open or SWs closes as a slave according to the status of D. 
     Operates  250  as a voltage step-up converter (voltage across Cf is lower than the desired output voltage across RL), SW remains closed over  255 . Ds prevents the degradation of the higher voltage across RL toward the lower voltage across Cf. SWs operates with a PWM, which determines over its duty cycle the output voltage across RL. The control difference value is taken as in the voltage step-down converter from the voltage value across Cf. This control difference value is transmitted together with a duty cycle information of SWs (its PWM signal or a digital signal corresponding to the PWM duty cycle of SWs) serially over the wireless coupling link to block  205 . 
       206  uses this data in a standard control function (I, PI or PID) and controls block  201 . In this manner the control loop is closed from RL over FSKLdata toward block  201 . Advantageously, this combined step-up respectively step-down configuration stabilizes the output voltage for both lower and higher received voltages on Cf. 
     It is also readily apparent that one or more  250  comprise one or more output voltages. 
     In another mode, the control difference value is derived from a plurality of output voltages, which are sampled in a time multiplex manner and/or be added through resistors. This control difference value is transmitted along with a duty cycle information of one or more switches SW (their PWM signals or a digital signal corresponding to the PWM duty ratios of a plurality of switches) serially over the wireless coupling link to block  205 .  206  uses this data in a standard control function (I, PI or PID) and controls block  201 . In this manner the control loop is closed from RL over FSKLdata toward block  201 . 
     If a plurality of  250  operate with one  200 ,  206  combines multiple control difference values of a plurality of received FSKLdata of coupled  250  and controls  201 . In this way, the total radiated power of  200  is adjusted to the overall power level required to supply a plurality of  250 . The fine output voltage regulation within the individual  250  is controlled by SW and/or SWs. 
     It is also possible to operate one or more  250  with continuously closed SW together with one or more  250 , which use SW and/or SWs as output voltage regulator on one  200 . 
     These output voltage control or regulation methods respectively their different modes correspond to the normal operating mode (continuous operation), while OCL is not active. For this case, the period comparison versus a predetermined reference value generates inside  250  a first result, which resets OCL to OFF (see  FIG.  4   ). This corresponds to the operating condition during the energy coupling condition characteristic for curve  1  in  FIGS.  1   c    and  1   d.    
     A further increase in the energy coupling (higher coupling k and/or larger load) results in critical- and finally overcritical coupling condition. The input impedance develops a plurality of different resonant frequencies (every zero crossing of the phase line of the input impedance). This causes in the control loop  202 ,  203 ,  203   a ,  203   b ,  206  and  207  a phase jitter around to the resonance frequency at critical coupling condition, which in excess leads to a superposition of two current or voltage curves at overcritical coupling condition. This state is defined here as over coupling and defines an unwanted operation status. The state change from critical toward overcritical coupling condition arises fluently in practice. 
     In practice, it is sufficient to detect the state of overcritical coupling explicitly for the energy coupling control of the wireless power transmission link. Depending on the implementation of the functions described in the following,  204  respectively  254  responds faster (even at critical coupling condition) or responds slower (not until overcritical coupling condition).  254  in  250  compares the half period of the detected zero-crossings versus a predetermined comparison value, which is a little less than the half period of the operation frequency in undercritical coupling condition. This comparison now yields a second result, which changes the state of OCL to ON (high potential). 
     In one embodiment, the period processor compares multiple of detected half periods versus almost the same value corresponding to the same multiple of half periods of the operating frequency in undercritical coupling condition. The response of the coupling detector  254  increases accordingly. 
     The ON in OCL forces  254  to generate an output signal, which opens SW as long as OCL is ON. The function of the voltage step-down converter is disabled and RL is decoupled from Cf. OCL remains ON even if the output of  252  indicating normal energy coupling conditions (detected half period is greater than the predetermined comparison value). 
     As a consequence, the lower load reduces the energy coupling. After an arbitrary time (TL) where  250  remains unloaded (RL is decoupled from Cf), the output of  254  changes independently (e.g. by means of an implemented one shot circuit or timer), OCL goes to OFF and RL restores coupling with Cf by closing SW (see  FIG.  4   ). The step-down converter restores normal operation. 
     When k and RL remain unchanged, the overcritical coupling condition reestablishes as soon as the system reaches the steady state condition in Lf and Cf. SW reopens via OCL for the time interval TL, then closes again etc. 
     Consequently, the coupling status of RL with Cf alters continuously. The average value of the effective RL appearing in the energy coupling reaches in this manner the value which is substantially equal to the boundary value of RL defining the critical coupling condition. This represents a limitation of the energy coupling to a maximum of one. Since  254  is responsive to one or a few half-periods and TL is much longer, the operation status during which the wireless power transmission link operates in the overcritical coupled condition reduces to a minimum or is prevented completely. Thereby, the entire power transmission link operates in the source- and load unit substantially continuously under resonant condition, i.e. at the resonance frequency. This resonance frequency is determined by  200  and  250 . 
     Advantageously this results in the smallest possible bandwidth. Additionally, the entire power transmission link operates always under matched conditions because always only the real part of the entire transmission path changes. A resonance frequency detuning is always readjusted in  200 . The effective real load resistance of  200  is determined by RL, the turn ratio of L 1  and L 2 , the window function (duty cycle) of the signal OCL and the energy coupling (k·Q). Thus, the wireless power transmission link directly corresponds to an RL (eventually transformed by L 1 /L 2 ) contacted at  200  and thus corresponds to a wired coupling. By choosing TL and selecting the components of the filter Lf and Cf, the ON/OFF timing of OCL can be designed flexibly. An abrupt coupling respectively abrupt decoupling of RL has no adverse impact on  200 , since Lf and Cf act as a filter, which smoothing any appearing load transients on  200 . 
     In  200  occurs something similar. If an overcoupling condition is detected in  204 , OCT sets to ON,  206  disables  201  or reduces the output power of  201  to a minimum value for a time interval TT. The time response of OCT is delayed by TD versus OCL. After the time interval TT has elapsed, the output of  204  changes independently (e.g. by means of an implemented one shot circuit or timer), OCT goes to OFF and  201  is again enabled or restored to normal output power (see  FIG.  4   ). If overcoupling condition is redetected, the process repeats:  201  again disables delayed by TD or reduces the output power and automatically reestablishes normal operation after the interval TT. In this manner a safe operation and a soft start at  200  is guaranteed whenever overcoupling condition occurs. 
     In one embodiment of the current invention,  204  is featured with a delayed time response versus the time response of  254  concerning the start of TD, for an appearing overcoupling condition. 
     In another embodiment,  204  and  254  are featured with an identical response time (same or similar implementation), then the required delay in the start of TD is implemented in  206  as a delay on the event OCT. Operationally, both versions are identical. TD ensures that the energy coupling control is always first executed at  250 . Advantageously, other  250 , which are coupled to  200  remain continuously supplied with energy due to the continuous operation of  200 . 
     TT within  200  and TL within  250  need not to be equal. The redundancy of the double implementation of  208  within  200  and  250  limit, in any case, the energy coupling (k·Q) substantially to the value one. This comprises also a possible malfunction in  250 : Then, the other block  208  reacts as soon as overcoupling condition has established. 
     A source unit can also radiate energy in a predetermined frequency range. Further, a load unit can drive a load as a regulated current source. This is illustrated in the block diagram in  FIG.  3   . The source unit  300  comprises a large signal Voltage Controlled Oscillator (VCO)  301 , which is powered with the necessary operating voltage by a controlled power source  302 . 
     The term large signal VCO stands for an oscillator whose active element  304  (amplifier) operates mainly as a switch. Further, the frequency of this oscillator is controlled by a current or a voltage. Advantageously, the frequency controller comprises at least one coupling switch, which couples at least one inductor or at least one capacitor via a variable coupling interval to the resonant circuit, wherein the coupling interval is smaller than a period of the resonant circuit. 
     The inverting amplifier  304  drives a series resonant circuit comprising the electrically controlled capacitor C 1  and the primary coil L 1  of the wireless power transmission link. An inverter  303  closes a feedback loop to the input of  304  and provides a continuous current- and voltage oscillation in the resonant circuit (positive feedback). 
     In a further embodiment of the present invention, the large signal VCO is equipped with an H-bridge or push-pull stage and is implemented with a parallel resonant circuit (not shown in  FIG.  3   ). The functions of all the properties described (Phase Locked Loop (PLL) operation and control loops) remain the same as for the large signal VCO driving the series resonant circuit. 
     The oscillator signal (output signal first of  303 ) is compared in a phase comparator  305  versus a reference value (fsoll). fsoll is generated in a frequency synthesizer  306  using a fundamental frequency (fref). The output signal from  305  is filtered by a loop filter  307  and subsequently coupled as a control factor to the variable capacitor C 1 . When  301  does not oscillate at fsoll,  305  generates an error output signal, which retunes C 1  after filtering with  307  until first becomes equal to fsoll. In this manner  300  emits electromagnetic energy, which is regulated in the frequency at fsoll. Any kind of frequency detuning such as by component tolerances, component aging or changes in the load unit  350  are corrected within a few dozen oscillator periods. The coupling detector  308  comprises the period processor  309 , because the zero-crossing- or sign detector essentially corresponds to  303 . An output signal of  305  is coupled to a data demodulator  311 . 
     Alternatively, in another embodiment,  311  uses first as its input signal.  311  operates substantially phase or frequency sensitive and couples demodulated data from the load unit (FSKLdata) to a function controller  310 . The output signal of  303  is coupled to a period processor  309 .  309  analyzes the input signals time- or frequency sensitive and generates in relation to a predetermined comparison value, an output signal (OCT), which in turn serves  310  as an input signal. An output signal of  310  controls  306  and defines in this manner fsoll. 
     For this purpose comprises  306  mainly a programmable frequency divider. In one embodiment of the actual invention the frequency divider within  306  changes its divisor as a function of time to form an arbitrary frequency spectrum (e.g. approximated rectangular spectrum or Sin(x)/x, etc.) in fsoll and accordingly in the radiated field. Another output of  310  controls the output voltage or the output current of  302 . 
     The secondary coil L 2  of the wireless power transmission link forms together with C 2  a parallel resonant circuit, which is tuned to fsoll (corresponding to the resonance frequency of C 1  and L 1 ). If fsoll is changed over time, the resonance frequency of L 2  and C 2  is selected so that it substantially matches the center of the frequency range, which in turn is radiated by  300 . 
     A rectifier  351   a ,  351   b  rectifies the resonance voltage and couples it through a low pass filter LF and a diode Dp to a filter capacitor Cf and the load RL. Parallel to  351   a  and  351   b  are each a switch SWa and SWb connected. Advantageously,  351   a  and SWa, respectively, and  351   b  and SWb are integrated into one component. If SWa and SWb are open, RL is coupled via  351   a  during one half resonant circuit period and via  351   b  during the other half resonant circuit period over Lf to the resonant circuit L 2 , C 2 . 
     If SWa and SWb are closed at the same time, the resonant circuit is shunted by SWa and SWb, Dp blocks and RL is no longer coupled with the resonant circuit L 2 , C 2 . The configuration SWa, SWb, C 2 , L 2 , Lf and Dp operate as a shunt regulator for RL. An analog- to digital converter  356  measures by means of a current sensor  356   a  the load current through RL.  355  generates responsive to the output signal of  356  a pulse width signal (PWM signal) and controls SWa and SWb. For this purpose  355  comprises at least: a voltage or current reference, a comparator, a loop filter and a PWM generator (not shown). A zero-crossing detector  352  signals a sign change in the resonance voltage. 
     Alternatively, in a further embodiment, the resonance current or the resonance voltage in the resonant circuit L 2 , C 2  is detected (not shown). The output signal from  352  synchronizes block  355  (the contained PWM generator), block  356  and the data modulator  353 . In addition, the output of  352  is also coupled to a period processor  354 .  354  corresponding to  309  analyzes the input signals time- or frequency sensitive and generates in relation to a predetermined comparison value, an output signal (OCL) that controls block  355 . 
     The output value of  356  is compared in  353  with a reference value (feedback versus reference comparison). A resulting control difference value is converted in  353  to an appropriate transmission format and along with other data, which are generated and/or stored in  353  coupled as FSKLdata to ModSWa and ModSWb. ModSWa and ModSWb modulate Cm with the load of  350 . FSKLdata is transmitted in this manner to  300 . 
     Alternatively, in a further embodiment, Cm is coupled at the tapping of L 2  via a ModSW to ground. 
     The output current control or regulation of the current invention can be carried out in various modes. The modes described below are individually and/or jointly active at different times as a combination in various embodiments.  355  signals the selected mode over the connection mode to  353 . This mode selection signal controls  353 , so that the relevant control data (control difference value) of the respective operation mode are transmitted. 
     In one mode, SWa and SWb remain for the output current regulation open. Thus,  356  detects the output current in RL. The control difference value (feedback to reference value comparison, i.e. comparison with a first reference value) in FSKLdata corresponds to the output current deviation from a desired output current through RL. This control difference value is serially transmitted via the wireless coupling link to block  311 .  310  uses this data in a standard control function (I, PI or PID) and controls block  302 . In this manner the control loop is closed from RL over FSKLdata toward block  302 . 
     In another mode, SWa and SWb operate as a PWM controller for the output current regulation in  350 .  356  detects the current through RL. The control difference value (feedback to reference value comparison, i.e. comparison with a second reference value) in FSKLdata corresponds to the current deviation from a desired current through RL. This control difference value is transmitted together with a duty cycle information of the switches SWa respectively SWb (its PWM signal or a digital signals corresponding to the PWM duty cycle of SWa respectively SWb) serially over the wireless coupling link to  311 .  310  uses this data in a standard control function (I, PI or PID) and controls block  302 . In this manner the control loop is closed from RL over FSKLdata toward block  302 . 
     This permits to optimize the efficiency of the wireless power transmission link and yet ensures the operation of several load units  350  at one source unit  300 . Additionally, the fast loop response (shortest loop) in the output current regulation by means of SWa respectively SWb results in the best dynamic properties. 
     It is also readily apparent that one or more  350  may comprise one or more output currents. 
     In another mode, the control difference value is derived from a plurality of output currents, which are sampled in a time multiplexed manner and/or are simply added. This control difference value is transmitted along with a duty cycle information of one or more switches SWa respectively SWb (their PWM signals or a digital signal corresponding to the PWM duty ratios of a plurality of switches) serially over the wireless coupling link to block  311 .  310  uses this data in a standard control function (I, PI or PID) and controls block  302 . In this manner the control loop is closed from RL over FSKLdata toward block  302 . 
     If a plurality of  350  operate with one  300 ,  310  combines multiple control difference values of a plurality of received FSKLdata of coupled  350  and controls  302 . In this way, the total radiated power of  300  is adjusted to the overall power level required to supply a plurality of  350 . The fine output current regulation within the individual  350  is controlled by SWa respectively SWb. 
     It is also possible to operate one or more  350  with continuously open SWa respectively SWb together with one or more  350 , which use SWa and SWb as output current regulator on one  200 . 
     These output current control methods respectively their different modes correspond to the normal operating mode (continuous operation), while OCL is not active. For this case, the period comparison versus a predetermined reference value generates inside  350  a first result, which resets OCL to OFF (see  FIG.  4   ). This corresponds to the operating condition during the energy coupling condition characteristic for curve  1  in  FIGS.  1   c  and  1   d   . A further increase in the energy coupling (higher coupling k and/or larger load) results in critical and finally overcritical coupling condition. The input impedance develops a plurality of different resonant frequencies (every zero crossing of the phase line of the input impedance). This causes in the control loop  301  phase jitter around the resonance frequency at critical coupling condition, which in excess leads to a superposition of two current or voltage curves at overcritical coupling condition. This state is defined here as over coupling and defines an unwanted operation status. The state change from critical-toward overcritical coupling condition arises fluently in practice. In practice, it is sufficient to detect the state of overcritical coupling explicitly for the energy coupling control of wireless power transmission link. Depending on the implementation of the functions described in the following,  309  respectively  354  responds faster (even at critical coupling condition) or responds slower (not until overcritical coupling condition).  354  in  350  compares the half period of the detected zero-crossings versus a predetermined comparison value, which is a little less than the half period of the operating frequency in undercritical coupling condition. This comparison now yields a second result, which changes the state of OCL to ON (high potential). 
     In one embodiment, the period processor compares multiple of detected half periods versus almost the same value corresponding to the same multiple of half periods of the operation frequency in undercritical coupling condition. The response of the coupling detector  354  increases accordingly. The ON in OCL forces  354  to generate an output signal, which closes SWa and SWb as long as OCL is ON. Consequently, RL is decoupled from C 2 , L 2  by Dp. OCL remains ON even if the output of  352  indicating normal energy coupling conditions (detected half period is greater than the predetermined comparison value). 
     As a consequence, the lower load reduces the energy coupling. After an arbitrary time (TL) where  350  remains unloaded (RL is decoupled from C 2 ), the output of  354  changes independently (e.g. by means of an implemented one-shot circuit or timer), OCL goes to OFF and RL restores coupling with C 2 , L 2  by opening SWa and SWb (see  FIG.  4   ). The rectifier  351   a ,  351   b  restores normal operation. When k and RL remain unchanged, the overcritical coupling condition reestablishes as soon as the system reaches the steady state condition in Lf and Cf. SWa and SWb close again via OCL for the time interval TL, then open again, etc. Consequently, the coupling status of RL with C 2 , L 2  alters continuously. The average value of the effective RL appearing in the energy coupling reaches in this manner the value which is substantially equal to the boundary value of RL defining the critical coupling condition. This represents a limitation of the energy coupling to a maximum of one. 
     Since  354  is responsive to one or a few half-periods and TL is much longer, the operation status during which the wireless power transmission link operates in the overcritical coupled condition reduces to a minimum or is prevented completely. Thereby, the entire power transmission link operates in the source- and load unit substantially continuously under resonant condition, i.e. at the resonance frequency. This resonance frequency is determined by  300  and  350 . Advantageously this results in the smallest possible bandwidth. Additionally, the entire power transmission link operates always under matched conditions because always only the real part of the transmission path changes. A resonance frequency detuning is always readjusted in  300 . The effective real load resistance of  300  is determined by RL, the turn ratio of L 1  and L 2 , the window function (duty cycle) of the signal OCL and the energy coupling (k·Q). 
     Thus, the wireless power transmission link directly corresponds to an RL (eventually transformed by L 1 /L 2 ) contacted at  300  and thus corresponds to a wired coupling. By choosing TL and selecting the components of the filter Lf and Cf, the ON/OFF timing of OCL can be designed flexibly. An abrupt coupling respectively abrupt decoupling of RL has no adverse impact on  300 , since Lf and Cf act as a filter, which smoothing any appearing load transients on  300 . 
     In  300  occurs something similar. If an overcoupling condition is detected in  309 , OCT sets to ON,  310  disables  302  or reduces the output power of  302  to a minimum value for a time interval TT.  350  is no longer supplied with energy. The time response of OCT is delayed by TD versus OCL. After the time interval TT has elapsed, the output  309  changes independently (e. g. by means of an implemented one shot circuit or timer), OCT goes to OFF and  302  is again enabled or restored to normal output power (see  FIG.  4   ). If overcoupling condition is redetected, the process repeats:  302  again disables delayed by TD or reduces the output power and automatically reestablishes normal operation after the interval TT. In this manner, a safe operation and a soft start at  300  is guaranteed whenever overcoupling condition occurs. 
     In one embodiment of the current invention,  309  is featured with a delayed time response versus the time response of  354  concerning the start of TD, for an appearing overcoupling condition. 
     In another embodiment,  309  and  354  are featured with an identical response time (same or similar implementation), then the required delay in the start of TD is implemented in  310  as a delay on the event OCT. Operationally, both versions are identical. TD ensures that the energy coupling control is always executed at  350 . Advantageously, other  350 , which are coupled to  300  remain continuously supplied with energy due to the continuous operation of  300 . 
     TT within  300  and TL within  350  need not to be equal. The redundancy of the double implementation of  308  within  300  and  350  limit, in any case, the energy coupling (k·Q) substantially to the value one. This comprises also a possible malfunction in  350 : Then, the other block  308  reacts as soon as overcoupling condition has established. 
     In one embodiment of the current invention,  310  changes the resonance frequency in  300  responsive to  306  and  305 . This represents another variant to reduce the energy coupling. In this manner, advantageously, the coupling k and/or the quality factor Q of  350  and above all the amplitude of  301  change. This may be sufficient to inhibit the overcoupling condition. Thus,  300  features an additional degree of freedom versus  200  by arbitrarily controlling the resonance frequency. This resonance frequency change occurs alone or combined with a control of  302 . 
     Advantageously, the described methods of the different modes of  200 ,  250 ,  300  and  350  of  FIGS.  2  and  3    serve greatest flexibility. Each  250  and  350  may determine itself the method by which the output voltage respectively the output current should be regulated. The control difference value, which is transmitted in FSKLdata describes a measure as to whether more or less power over the wireless transmission link from  200  respectively  300  is to be pushed to  250  respectively  350 . 
     In another embodiment of the current invention are, in the event of overcoupling condition, only individual partial loads, i.e. a number of output voltages of a plurality of output voltages in  250  respectively a number of output currents of a plurality of output currents in  350 , successively decoupled via a plurality of switches (SW), or diodes (DP) within  250  respectively  350 . In this manner, other partial loads of  250  respectively  350  remain continuously supplied with energy. 
     In general, the energy coupling limitation of the current invention limits the load energy or power. Depending on the implementation of the functions and operating modes described, an output voltage across RL and respectively an output current through RL may more or less drop. Thanks to the existing energy storages (Ls, Cs within  250  respectively Lf, Cf within  350 ), the output voltage respectively the output current might not interrupt necessarily. All modes of the described methods may interoperate with each other and the pairing of any one or more  250  respectively  350  with one  200  respectively  300  can be permuted arbitrarily. Advantageously, TL respectively TT is elected by factors greater than the lock time of the PLL loop  202 ,  203 ,  203   a ,  203   b ,  206 , and  207  respectively  301 ,  305  and  307 . 
       FIG.  5   a    shows a typical flexible implementation according to one embodiment of the current invention: A transmitter and/or receiver coupling coil (LNFC) is used in two different frequency ranges at the same time and/or at different times in two different main applications. 
     The first main application is an RFID or Near Field Communication (NFC) by inductively coupled inductors and operates in a first frequency range above 1 MHz (e.g. 6.78 MHz or 13.56 MHz). The capacitors CNFC 1  and CNFC 2  tune LNFC to the first frequency range and/or match LNFC to the input or output of the processing unit  500  of the first main application. In addition, CNFC 1  and CNFC 2  decouple an operating status of the first main application from an operating status such as transmission or reception of the second main application. 
     Additionally, in a further embodiment of the invention, switches or amplitude limitation means (e.g. diodes) are implemented to protect the input or output stage within  500  from overvoltage as depicted in  FIGS.  5   b - 5   h   . E.g. limiting diodes (D) connected to a reference potential are used for that purpose. The switches (SW 1 , SW 2 ) are in series (SW 1 ) and/or implemented as switches (SW 2 ) to a reference potential. 
     The second main application is a wireless power transmission, and operates in a second frequency band below 1 MHz (for example, 120 . . . 135 kHz). The coils Lx and Ly tune LNFC to the second frequency range and/or match LNFC to the input or output of the processing unit  200 / 250 / 300 / 350  of the second main application. In addition, Lx and Ly decouple an operating status such as transmission or reception of the second main application from an operating status of the first main application. The processing unit  200 / 300 / 250 / 350  corresponds to one or multiple blocks in  FIG.  2    respectively  FIG.  3   , wherein multiple blocks can be operational at different times. In one embodiment of the actual invention the second main application operates as a transmitter and emits power by means of LNFC. 
     In another embodiment of the current invention, the second main application operates as a receiver and receives power by means of LNFC. Advantageously,  250  operates in the second main application as a voltage step-up converter, since the induced voltage due to the low inductance of LNFC is relatively small (in the range 4 uH . . . 20 uH) and the decoupling inductances Lx and Ly are rather large (in the range 30 . . . 100 uH). 
     If a plurality of  250  or  350  operate on one  200  or  300 , the communication of the individual FSKLdata may be disturbed. For this reason, the individual FSKLdata are transmitted in periodic- or random time division multiplex. As a consequence, ModSW remains open or closed if no FSKLdata are transmitted. 
       FIG.  6   a    shows two at different times transmitted FSKLdata 1  and FSKLdata 2 , whereby these data might be generated from the same or two different  250  or  350 . During the time intervals Tbusy no further FSKLdata can be sent because the channel is busy (the channel busy indicator FSKLdatajam is active).  253  and  353  comprise for this purpose a communication detector  601  (see  FIG.  6   b   ), which signals the time interval Tbusy. The input signal of block  601  is served by synch. Synch is analyzed within  601  in a time and/or frequency sensitive manner and generates at its output a binary signal COMready, which is OFF during the time interval Tbusy, otherwise set to ON. 
     The communication trigger  602  generates a positive output pulse COM in a periodic or random manner having the length of FSKLdata, wherein the positive output pulse COM is only generated during Tvalid. This positive output pulse COM signals the communication controller  603  to generate a serial bitstream of FSKLdata (or at least portions thereof) by means of the clock COMclk and couples FSKLdata to ModSW. COMclk is derived from synch by a frequency divider  604 . In this manner, advantageously, FSKLdata are always sent synchronously with respect to the resonant circuit periods. This simplifies  205  and  311  and provides good reception sensitivity. 
     FSKLdata transmits binary information, which is differentially encoded in two phases at a bitrate whose clock is divided from synch. The binary information is formatted in byte length. The communication protocol FSKLdata comprises data formatted in a header (constant bit length), the message (variable bit length), and a checksum (constant bit length). 
     The header defines the message portion of the following message and its length in bytes. E.g. the lower 4 bits of the header defines the message portion and the upper 4 bits defines the message length (see  FIG.  7   ). In this case, one or more message portions can vary in their length (e.g. control variable in  FIG.  7   f   ). The message transmits the message portion, which was previously signaled in the header. The checksum transmits a CRC- or hash value calculated over the message portion or alternatively also calculated over the message including the header. 
     In a first message portion at least one identification number which is pre-stored in  250  or  350  is transmitted (see  FIG.  7   a   ). 
     In a further embodiment of the current invention, an identification number is generated for a time interval (e.g. duration of energy transmission session). 
     In another message portion, an identification number is transmitted (see  FIG.  7   b   ), which characterizes a communication protocol version of the message (header/message portion encoding according to the pattern as shown in  FIG.  7    or similar). 
     In another message portion, at least one value characterizing power or characterizing a power class is transmitted (see  FIG.  7   c   ). 
     In a further message portion, at least one value of a secondary receiver inductor (L 2 ) and/or the number of windings of L 2  is transmitted (see  FIG.  7   d   ). 
     In a further message portion, at least one value of a resonance frequency and/or a reception bandwidth is transmitted (see  FIG.  7   e   ). 
     In another message portion, at least a control difference value or a control trend is transmitted (see  FIG.  7   f   ). 
     In another message portion, at least a value of a coupling status or an energy coupling status (OCL flag) and/or a operation status is transmitted (see  FIG.  7   g   ). The OCL flag is a bit that is set by the OCL signal at  253  respectively  353 . 
     In another message portion, at least a value of a not allowed frequency range (i.e. an excluded transmission frequency or bandwidth) is transmitted (see  FIG.  7   h   ). 
       200  or  300  reads these values and adjusts operation mode conditions according to the received message or message portion. 
     Thus, in one embodiment of the current invention, at least one or more message portions of the received message are processed according to the received protocol identification. 
     In another embodiment,  201  respectively  302  is controlled in the output power responsive to the control difference value, and if necessary limited in the output power responsive to the power class and/or to the inductor L 2  and/or to the number of turns of L 2 . E.g. if a control difference value reaches a predetermined value, then the radiated power is limited or reduced until the received control difference value does no longer indicate the predetermined value. This guarantees overdrive protection of all  250  and  350 . 
     Additionally, as an option,  201  respectively  302  is enabled respectively disabled (continuous energy transmission respectively no energy transmission) or continuously switched between these two operation states (burst mode) responsive to the received operation status and/or the received energy coupling status information (OCL flag) in FSKLdata. 
     In a further embodiment,  306  is controlled in fsoll such that the radiated power at a predetermined resonance frequency or over a frequency bandwidth, under consideration of a not allowed frequency or frequency range, is in accordance with a predetermined value or at least approximates the predetermined value. 
     In a further embodiment,  310  is controlled such that the emitted electromagnetic power is disabled or at least the emitted power is limited within a frequency range responsive to a specific identification numbers. In this manner, the power flow to individual  250  or  350  is controlled, or the power flow is enabled or disabled responsive to an identification number. 
       FIG.  8    shows an example of a detailed circuit according to  300  in  FIG.  3   .  FIGS.  9   a  and  9   b    show signal waveforms, which are referred to in the further description. A large signal VCO, here a parallel push-pull circuit with Q 1  and Q 2  drives the parallel resonant circuit comprising L 1 , C 1 . DR energizes the center tap of L 1  by a controllable power supply SMPS. Advantageously, the voltage across L 1  in this resonant circuit topology is independent of the quality of the resonant circuit and is further proportional by a constant to the supply voltage at the output of the SMPS. R 8 , D 3  and D 4  switch Q 5  always conductive, except during the voltage zero crossings of the resonant circuit voltage. 
     The voltage across R 7  clocks the D flip-flop FF 1 , which operates as a frequency divider. The output signals are delayed by N 1  . . . N 4  and further coupled in an AC manner via CK 1 , CK 2 , D 1 , D 2 , R 5  and R 6  to Q 6  . . . Q 9 . Q 6  . . . Q 9  drive Q 1  and Q 2  by means of R 1  and R 2 . The output signal first of Q 5  is also coupled to the input of the phase comparator PFD of block  801 , which compares first versus fsoll.  801  corresponds substantially to  305  and  307 . 
     The charge pump R 9 , R 10 , Q 14 , Q 15 , Q 16  and Q 17  generates in the PLL loop filter Ci, Cp and Rp, an error voltage corresponding to the phase difference between first and fsoll. This error voltage generates with the current in Q 18  . . . Q 20  a ramp voltage across Cr, that is modified by repeating short conducted Q 21  to a sawtooth voltage (waveform C). First is formed by NOR and N 6  into short time pulses now named as synch that controls Q 21  and synchronizes the FSKLdata demodulation in the controller (signal curve synch). 
     This sawtooth voltage alters the state of N 5 , as soon as its threshold voltage is reached (see  FIG.  9   b    T 1 , T 3 , T 4 , T 6  intersections with the dotted line curve C and output D). The following OR  1  and OR 2  route the timing signal D to the corresponding coupling switches Q 3  respectively Q 4  to control the resulting value of capacitor C 1  in a large signal manner. The driving signals of FF 1  are inverse to each other and change their state at  0 , T 3 , and T 6 . In this manner, only every second period of D is routed to E respectively F. The drivers Q 10  . . . Q 13  switch Q 3  respectively Q 4  accordingly via R 3  and R 4 . As a consequence, across Q 3  respectively Q 4  develops alternately a half sine wave per period (see waveform VQ 3  dashed respectively VQ 4 ). R 1  . . . R 4  have a low value and prevent transients in the switch control. Q 3  and Q 4  closing during the interval T 2 , T 3  (Q 4 ) and the interval T 5 , T 6  (Q 3 ) independently through their internal diodes as soon as C 1 ′ respectively C 1 ″ has discharged. 
     This phase control of the effective parallel capacitance of C 1 ′ and C 1 ″ provides a control range of C 1 tot=C 1 +0.5·C 1 ′, if C 1 ′=C 1 ″. Fist is in this manner always perfectly adjusted to fsoll. Fref is divided by the programmable divider synth to obtain fsoll. Delays N 1  . . . N 4  compensate the propagation delay of fist to D and ensure the full control range in C 1 tot. One output of FF 1  is coupled to the clock inputs of FF 2  and FF 3 . FF 2  is set with the rising edge and after reaching the threshold over CT 1  reset. RT 1  and CT 1  define the time constant (predetermined reference value). 
     If signal curve v reaches the threshold level of the reset input of FF 2 , then only a “OFF” is latched to the output of FF 3  (see waveform u). If the threshold level of the reset input of FF 2  is not reached (because drive operates at a too high frequency), then a “ON” is latched to the output of FF 3 . This situation occurs after the second drive period in  FIG.  9   a   , since drive in this example has such a high frequency that v never reaches the threshold (continuous over-coupling). 
     D 5  ensures the reset of the voltage across CT 1 . The mono stable flip-flop FF 4 , RTT and CTT generate a high (ON) in OCT when u remains set for more than one drive period. After a time interval (TT) has elapsed, the voltage across CTT reaches the threshold, which resets FF 4 . OCT is coupled to the controller, if necessary, internally in the controller additionally delayed, and then used over PBUS to control the SMPS. A signal responsive to OCT is signaling the energy coupling status visually by means of LED D 7  (over coupling indicator). 
     The controller comprises the divider synth and also the FSKLdata demodulator. In this way, the frequency deviation is measured by a cycle counter over a temporal profile. The received FSKLdata control the SMPS and/or synth, and optionally further optical indicators (not shown in  FIG.  8   ). For this purpose, the controller comprises a microcontroller with RAM and ROM memory and/or a PLD/FPGA and/or ASIC components. These implementation methods are known to a person skilled in the art. 
       FIG.  10    shows the detailed circuit of a receiver  250  of a wireless power transmission system according to  FIG.  2   . L 2  form together with C 2  a series resonant circuit for receiving the electromagnetic field lines. GL rectifies the received signal and couples it after filtering with Lf and Cf to SW within  1000 . This block corresponds to a voltage step-down converter and represents a typical implementation of LM 5116  by an excerpt of its data sheet. D is implemented as a switch and LS, Cs are storage elements. The exact detail function of LM 5116  and its components shown here is referred to the data sheet. The operation mode of LM 5116  is carried out in diode emulation mode, or alternatively in the synchronous mode. Rfb 1  and Rfb 2  couple the output voltage across RL in a voltage divider manner to the feedback input FB. The switching frequency of the PWM generator inside LM 5116  is here directly synchronized with the electromagnetic field. For this purpose, R 8 , D 3  and D 4  keep Q 5  conducting, except during the minima of the rectified resonant circuit voltage. N 6  and NOR form the voltage across R 7  into short pulses, which control over Csync and RT the RT input of LM 5116 . This synchronizes the internal PWM and switching frequency of  1000  with the received electromagnetic alternating field received at L 2 , C 2 . Advantageously,  1000  discharges the resonant circuit L 2 , C 2  (filtered by LF, Cf) on a resonant circuit half-period base and the energy charging respectively the energy discharge of the resonant circuit L 2 , C 2  and its load substantially coincides. This reduces the required capacitor size and currents in Cf versus the unsynchronized case with the same filter properties. As a consequence, the cost reduces and the reliability increases. 
     In addition, the NOR output (synch) is coupled to the synchronization input of the Controller/AD. The voltage across R 7  is frequency divided in FF 1  and thereafter analyzed in  1001  whether overcoupling condition has developed or not.  1001  corresponds to  802 , and hence the waveforms of  FIG.  9    are valid. The output signal of  1001  (OCL) is inverted by Nen and coupled to the EN input of LM 5116 . If overcoupling condition is detected (OCL is at high level), thus SW opens and the voltage down converter interrupts the energy flow from Cf to RL. 
     OCL is processed together with other values that shall be transmitted within the Controller. Here refer to OCT and Controller in  FIG.  8   . The Controller shown in  FIG.  8    establishes an additional time response delay for the operational behavior with the load unit in  FIG.  10   . By means of this delay, the resulting OCT appears always after OCL (see  FIG.  4   ). The timing of OCT and OCL in the overall wireless power transmission procedure is indicated in  FIG.  4   . 
     The timing shown in  FIG.  9   a    represents a quantitative illustration of  1001 . 
     In another embodiment of the current invention,  1001  is eliminated in  FIG.  10   . Here, the signal OCL is directly determined in the Controller/AD by counting fref clock cycles within one synch period. For this case, the Controller according to  FIG.  8    can be featured with a smaller or no additional delay. 
     In  FIG.  10   , the output voltage across RL, and the voltage across Cf is coupled by means of an additional voltage divider RF 1  and RF 2  to the Controller/AD. The existing analog-digital converter (AD) within Controller converts these input values into a digital format. 
     These data are coupled in the above described manner (protocol formation and redundancy addition), and after adding other values (see  FIG.  7   ) as FSKLdata via the drivers and R 4  to ModSW. ModSW sets Cm while numbers of resonant circuit periods corresponding to the bit rate, to ground, and modifies in this manner the capacitance of C 2 . 
     The bit rate is obtained from down divided synch in the Controller/AD of  FIG.  6   b   . In the case the Controller/AD is implemented as sequential program (software), then the implementation of  FIG.  6   b    is slightly different. It is important that the data rate (bit rate) is synchronized with synch and a possible data transfer conflict (FSKLdatajam) is avoided. For this purpose, the Controller/AD detects period- or frequency changes of the signal synch and any received FSKLdata of other load units by means of reference frequency. 
     Alternatively, in an embodiment of the present invention any other FSKLdata are demodulated (e.g. at least a portion of  FIG.  7   ) and shown to the user. The Controller/AD requires beside data processing means internal ND converters, RAM and ROM memory, and their coupling means (MBUS) to the controller. The processing frequency (clock) is fref or alternatively multiple factors of synch. R 6  and C 3  eliminate or reduce voltage transients at the output of GL (snubber). The resulting energy is dissipated across a resistor, or alternatively, as shown here, used as an additional voltage V+(e.g. for the Controller/AD). For this purpose D 2  charges C 4  and D 1  limits V+. Alternatively, a coil is coupled in series to D 2  (not shown). In this manner, the load unit comprises always a minor base load. Further, this base load is also determined by Ruv 1 , Ruv 2 , Rf 1 , Rf  2  and the base load of LM 5116 . 
     It is obvious for a person skilled in the art that various described methods and/or modes of  FIGS.  2 ,  3 ,  8  and  10    comprise or include digital signal processing. This is implemented in a discrete or integrated manner, as a programmable logic device (PLD, FPGA), and/or as piece of hardware or software within a microcontroller. Furthermore various numbers of sub-blocks within the  FIGS.  2 ,  3 ,  8  and  10    can be integrated in integrated circuits (ASICs). 
     In one embodiment of the current invention, synch is served as a reference frequency for other communication means, such as e.g. NFC, RFID, Bluetooth, WLAN, UWB, etc. (see output synchout in  FIG.  10   ). For this purpose synch is multiplied by means of a PLL to an integer number, and thereafter used to control a data rate (bit rate) or to define and synchronize the transmission frequency and/or a communication time window (communication time slot) of a time multiplex transmission and/or to control a random code sequence of a code division multiplex system (frequency hopping FH- or a direct sequence method).