Patent Publication Number: US-2023142184-A1

Title: Method, System, and Apparatus for Resonator Circuits and Modulating Resonators

Description:
CROSS REFERENCE TO RELATED APPLICATIONS - CLAIMS OF PRIORITY 
     This application is a continuation of co-pending U.S. Application No. 17/032,694 filed Sep. 25, 2020, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS”, to issue on Oct. 18, 2022 as U.S. Pat. No. 11,476,823; and Application no. 17/032,694 is a continuation of U.S. Application No. 16/453,409 filed Jun. 26, 2019, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS”, now U.S. Pat. No. 10,790,796, issued Sep. 29, 2020; and Application No. 16/453,409 is a continuation of U.S. Application No. 15/607,388 filed May 26, 2017, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS”, now U.S. Pat. No. 10,355,663 issued Jul. 16, 2019; and Application No. 15/607,388 is a continuation of U.S. Application No. 15/046,363 filed Feb. 17, 2016, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS”, now U.S. Pat. No. 9,698,752 issued Jul. 4, 2017; and Application Number 15/046, 363 is a divisional of U.S. Application No. 14/214,119 filed Mar. 14, 2014, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS”, now U.S. Pat. No. 9,300,038 issued Mar. 29, 2016; and Application Number 14/214,119 claims priority under 35 USC 119 to U.S. Provisional Application No. 61/801,699 filed Mar. 15, 2013, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-8), and Application Number 14/214,119 is a Continuation-in-part (CIP) of commonly assigned and co-pending U.S. Utility Application No.13/316,243 filed Dec. 9, 2011, now U.S. Pat. No. 9,041,484 issued on May 26, 2015, and entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PAP), which 13/316,243 application claims priority under 35 USC 119 to the following U.S. Provisional Pat. Applications: Provisional Application No. 61/422,009 filed Dec. 10, 2010 and entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV), U.S. Provisional Application No.61/438,204 filed Jan. 31, 2011, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-2), U.S. Provisional Application No. 61/497,819 filed Jun. 16, 2011, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-3), U.S. Provisional Application No. 61/521,590 filed Aug. 9, 2011, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-4), U.S. Provisional Application No. 61/542,783 filed Oct. 3, 2011, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-5), and U.S. Provisional Application No. 61/565,413 filed Nov. 30, 2011, entitled “METHOD, SYSTEM, AND APPARATUS FOR RESONATOR CIRCUITS AND MODULATING RESONATORS” (attorney docket number PER-060-PROV-6); and the contents of each application and patent cited above are hereby incorporated herein by reference as if set forth in full. 
    
    
     TECHNICAL FIELD 
     Various embodiments described herein relate generally to resonator circuits and modulating resonators, including systems, apparatus, and methods employing resonators. 
     BACKGROUND INFORMATION 
     It may be desirable to modulate one or more resonators including shifting its resonate and anti-resonate points and provide resonator circuits, the present invention provides such modulation and circuits. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG.  1 A  is a simplified block diagram of duplex signal transceiver architecture according to various embodiments. 
         FIG.  1 B  is a simplified diagram of an RF channel configuration according to various embodiments. 
         FIG.  1 C  is a simplified, partial diagram of a section of the RF channel configuration shown in  FIG.  1 B . 
         FIG.  1 D  is a simplified, partial diagram of a section of the RF channel configuration shown in  FIG.  1 B  with a filter characteristic applied to a first band according to various embodiments. 
         FIG.  1 E  is a simplified, partial diagram of a section of the RF channel configuration shown in  FIG.  1 B  with a filter characteristic applied to a second band according to various embodiments. 
         FIG.  1 F  is a simplified, partial diagram of a section of the RF channel configuration shown in  FIG.  1 B  with a filter characteristic applied to a subunit or band of a first band according to various embodiments. 
         FIG.  1 G  is a simplified, partial diagram of a section of the RF channel configuration shown in  FIG.  1 B  with a filter characteristic applied to a subunit or band of a second band according to various embodiments. 
         FIG.  2 A  is a block diagram of an electrical signal filter module including resonators according to various embodiments. 
         FIG.  2 B  is a block diagram of a filter module including electrical elements representing the characteristics of a resonator according to various embodiments. 
         FIGS.  2 C and  2 I  are block diagrams of modulated or tunable resonator modules according to various embodiments. 
         FIGS.  2 D-H and  2 J  are block diagrams of tunable filter modules including tunable or modulated resonators according to various embodiments. 
         FIGS.  3 A- 3 C  are diagrams of capacitor modules that may be coupled to AW according to various embodiments. 
         FIG.  3 D  is a diagram of a tunable capacitor module that may be coupled to AW according to various embodiments. 
         FIG.  3 E  is a diagram of a tunable capacitor module that may be coupled to AW according to various embodiments. 
         FIG.  4    is a block diagram of fabrication configuration for a tunable filter module including tunable resonators according to various embodiments. 
         FIG.  5 A  is a block diagram of an electrical signal filter module including switchable resonators according to various embodiments. 
         FIGS.  5 B- 5 D  are block diagrams of switchable resonator modules according to various embodiments. 
         FIGS.  5 E- 5 F  are block diagrams of tunable, switchable filter modules including tunable or modulated, switchable resonators according to various embodiments. 
         FIG.  5 G  is a block diagram of a tunable, switchable filter module including tunable or modulated resonators according to various embodiments. 
         FIGS.  6 A- 6 F  are diagrams of filter responses of tunable, switchable filter modules according to various embodiments. 
         FIG.  7 A  is a block diagram of a filter module according to various embodiments. 
         FIG.  7 B  is a block diagram of a filter module including resonators according to various embodiments. 
         FIG.  8 A  is a block diagram of a switchable filter module according to various embodiments. 
         FIG.  8 B  is a block diagram of a switchable filter module including resonators according to various embodiments. 
         FIG.  8 C  is a block diagram of a tunable, switchable filter module including tunable or modulated resonators according to various embodiments. 
         FIG.  9 A  is a block diagram of a filter module according to various embodiments. 
         FIGS.  9 B- 9 C  are block diagrams of a tunable, switchable filter module including tunable or modulated resonators according to various embodiments. 
         FIGS.  10 A- 10 B  are diagrams of filter responses of tunable, switchable filter modules according to various embodiments. 
         FIG.  11    is a diagram of a filter frequency response according to various embodiments. 
         FIG.  12    is a flow diagram of a filter response selection method according to various embodiments. 
         FIG.  13 A  is a simplified block diagram of a filtering architecture according to various embodiments. 
         FIG.  13 B  is a block diagram of a filter architecture including modulated or tunable resonator modules and a resonator module according to various embodiments. 
         FIG.  14 A  is a diagram of a filter frequency response of a resonator module according to various embodiments. 
         FIG.  14 B  is a diagram of a filter frequency response of a modulated or tunable resonator module according to various embodiments. 
         FIG.  14 C  is a diagram of a filter frequency response of a filter architecture including a modulated or tunable resonator module and a resonator module according to various embodiments. 
         FIG.  15 A  is a simplified diagram of an RF channel configuration according to various embodiments. 
         FIG.  15 B  is a simplified diagram of an RF channel configuration with a channel in a first mode according to various embodiments. 
         FIG.  15 C  is a simplified diagram of an RF channel configuration with a channel in a second mode according to various embodiments. 
         FIG.  16 A  is a simplified block diagram of a filtering architecture according to various embodiments. 
         FIG.  16 B  is a block diagram of a filter architecture including switchable, modulated or tunable resonator modules and a resonator module according to various embodiments. 
         FIG.  16 C  is a block diagram of another filter architecture including switchable, modulated or tunable resonator modules and a resonator module according to various embodiments. 
         FIG.  16 D  is a block diagram of another filter architecture including switchable, modulated or tunable resonator modules and a resonator module according to various embodiments. 
         FIG.  16 E  is a simplified block diagram of signal transceiver architecture according to various embodiments. 
         FIG.  17 A  is a diagram of a filter frequency response of a resonator module according to various embodiments. 
         FIG.  17 B  is a diagram of a filter frequency response of a switchable, modulated or tunable resonator module in a first mode according to various embodiments. 
         FIG.  17 C  is a diagram of a filter frequency response of a filter architecture including a switchable, modulated or tunable resonator module in a first mode and a resonator module according to various embodiments. 
         FIG.  17 D  is a diagram of a filter frequency response of a resonator module according to various embodiments. 
         FIG.  17 E  is a diagram of a filter frequency response of a switchable, modulated or tunable resonator module in a second mode according to various embodiments. 
         FIG.  17 F  is a diagram of a filter frequency response of a filter architecture including a switchable, modulated or tunable resonator module in a second mode and a resonator module according to various embodiments. 
         FIG.  17 G  is a diagram of a filter frequency response of a filter architecture including a first switchable, modulated or tunable resonator module, a first resonator module, a second switchable, modulated or tunable resonator module, and a second resonator module according to various embodiments. 
         FIG.  18    is a flow diagram of a combined filter configuration method according to various embodiments. 
         FIG.  19 A  is a block diagram of an electrical signal filter module including resonators and diagrams of filter frequency responses of resonators according to various embodiments. 
         FIG.  19 B  is a diagram of filter frequency responses of the electrical signal filter module including resonators of  FIG.  19 A  in a first, pass-band filter mode according to various embodiments. 
         FIG.  19 C  is a diagram of filter frequency responses of the electrical signal filter module including resonators of  FIG.  19 A  in a second, notch filter mode according to various embodiments. 
         FIG.  19 D  is a diagram of combined filter frequency responses of the electrical signal filter module including resonators of  FIG.  19 A  in the first, pass-band filter mode according to various embodiments. 
         FIG.  19 E  is a diagram of combined filter frequency responses of the electrical signal filter module including resonators of  FIG.  19 A  in a second, notch filter mode according to various embodiments. 
         FIG.  20 A  is a block diagram of a tunable filter module including electrical elements representing the characteristics of tunable resonators according to various embodiments. 
         FIG.  20 B  is a block diagram of another tunable filter module including electrical elements representing the characteristics of tunable resonators according to various embodiments. 
         FIG.  21 A  is a block diagram of an electrical signal filter module including resonators and diagrams of filter frequency responses of resonators according to various embodiments. 
         FIG.  21 B  is a diagram of filter frequency responses of the electrical signal filter module including resonators of  FIG.  21 A  in a notch filter mode according to various embodiments. 
         FIG.  21 C  is a diagram of combined filter frequency responses of the electrical signal filter module including resonators of  FIG.  21 A  in the notch filter mode according to various embodiments. 
         FIG.  22 A  is a diagram of a resonant frequency probably function representing manufacturing variations for an acoustic wave (AW) device the according to various embodiments. 
         FIG.  22 B  is a diagram of an anti-resonant frequency probably function representing manufacturing variations for an acoustic wave (AW) device the according to various embodiments. 
         FIG.  22 C  is a diagram of a resonant frequency function representing temperature variations for an acoustic wave (AW) module the according to various embodiments. 
         FIG.  22 D  is a diagram of a capacitance per unit area probably function representing manufacturing variations for a capacitor module the according to various embodiments. 
         FIG.  23    is a block diagram of a configuration for a tunable filter module including tunable resonators according to various embodiments. 
         FIG.  24    is a flow diagram of a component modeling, manufacturing, and configuration method according to various embodiments. 
         FIG.  25 A  is a simplified block diagram of a signal filter architecture according to various embodiments. 
         FIG.  25 B  is a simplified block diagram of a signal filter architecture according to various embodiments. 
         FIGS.  26 A to  27 C  are diagrams of filter frequency responses of a signal filter architecture according to various embodiments. 
         FIG.  28 A  is a simplified block diagram of a signal filter architecture according to various embodiments. 
         FIG.  28 B  is a simplified block diagram of a signal filter architecture according to various embodiments. 
         FIGS.  29 A and  29 B  are diagrams of filter frequency responses of a signal filter according to various embodiments. 
         FIG.  30 A  is a simplified block diagram of a signal filter architecture according to various embodiments. 
         FIG.  30 B  is a simplified block diagram of an impedance matched (“IM”) signal filter architecture according to various embodiments. 
         FIG.  30 C  is a simplified block diagram of an impedance matched (“IMM”) signal filter architecture including an IM module according to various embodiments. 
         FIG.  30 D  is a simplified block diagram of IM signal filter architecture including an IMM, the IMM including an acoustic wave module (AWM) according to various embodiments. 
         FIG.  30 E  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 F  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 G  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing effective components according to various embodiments. 
         FIG.  30 H  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing redistributed effective components according to various embodiments. 
         FIG.  30 I  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 J  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing effective components according to various embodiments. 
         FIG.  30 K  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing redistributed effective components according to various embodiments. 
         FIG.  30 L  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 M  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing effective components according to various embodiments. 
         FIG.  30 N  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing redistributed effective components according to various embodiments. 
         FIG.  30 O  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 P  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing effective components according to various embodiments. 
         FIG.  30 Q  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing redistributed effective components according to various embodiments. 
         FIG.  30 R  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 S  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing effective components according to various embodiments. 
         FIG.  30 T  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) showing redistributed effective components according to various embodiments. 
         FIG.  30 U  is a simplified block diagram of IM signal filter architecture including an IMM, the IMM including an acoustic wave module (AWM) coupled to an adjustable capacitor according to various embodiments. 
         FIG.  30 V  is a simplified block diagram of IM architecture including an acoustic wave module (AWM) and matching components module (MCM) according to various embodiments. 
         FIG.  30 W  is a simplified block diagram of IM architecture including acoustic wave modules (AWM) acting as a matching components module (MCM) according to various embodiments. 
         FIGS.  31 A-C  are diagrams of frequency responses for various modules according to various embodiments. 
     
    
    
     DETAILED DESCRIPTION 
       FIG.  1 A  is a simplified block diagram of duplex signal transceiver architecture  10  according to various embodiments. As shown in  FIG.  1 A , architecture  10  includes a power amplifier module (PA)  12 , signal duplexer module  20 , radio frequency (RF) switch module  40 , low noise amplifier (LNA) module  14 , mixer module  60 A, and RF signal antenna  50 . In operation a signal  8  to be transmitted on the antenna  50  may be amplified via the PA module  12 , filtered by the duplexer module  20 , and coupled to the antenna  50  via the RF switch module  40 . In a duplex signal architecture a received signal on the antenna  50  may be simultaneously processed the duplexer module  20 . The resultant receive signal  24  may be amplified by the LNA module  14  and down-mixed to a baseband signal  60 C via the mixer module  60 A and a reference frequency signal  60 B. 
       FIG.  1 B  is a simplified diagram of an RF channel configuration  70 A according to various embodiments. As shown in  FIG.  1 B , a transmit (TX) band  73 A and a receive (RX) band  73 B may be located in close frequency proximity. The TX band may have a width defined by  72 A,  72 B (start and end of the TX band), the RX band may have a width defined by  72 C,  72 D (start and end of the RX band), and the frequency separation between the bands may be the difference between  72 C and  72 B (start of the RX band and end of the TX band). The TX band  73 A and the RX band  73 B may include a plurality of sub-bands or units  74 A,  74 B,  74 C and  75 A,  75 B, and  75 C as shown in  FIGS.  1 C to  1 G . 
     At the antenna  50  the TX band signal energy  73 A may be greater than the RX band signal energy as shown in  FIGS.  1 B to  1 G . Such a differential in signal energy may saturate the LNA module  14  and occlude the RX signal  24  in duplexed signal architecture  10 . The duplexer module  20  may include one or more filters (shown in  FIG.  2 F ) to limit interference of TX and RX signals in the TX and RX bands  73 A,  73 B. The combined TX and RX signal  42  may be communicated according to one or more communication protocol or standards including Code Division Multiple Access (“CDMA”), Wide Band Code Division Multiple Access (“W-CDMA”), Worldwide Interoperability for Microwave Access (“WIMAX”), Global System for Mobile Communications (“GSM”), Enhanced Data Rates for GSM Evolution (EDGE), and other radio communication standards or protocols. Such standards or protocols may provide minimum signal separation or interference mitigation requirements for communication of signals on the respective networks via an antenna  50 . 
     The PA module  12  may also introduce noise or interface due to its fall off in power about the TX band to be amplified. The excess PA power may interfere with the LNA module  14  operation. A blocker signal near the TX, RX bands  73 A,  73 B or between same present on the antenna  50  (may be due to other signals in the communication network) may also interfere with the LNA module  14  operation and cause loss in the RX signal  24 . 
     Duplex systems or architecture  10  may employ filter modules including duplexer modules. The duplexer modules may include known filter elements such as resistors, capacitors, inductors, digital signal processors (DSPs), and resonators. Configurations of these components may form filter modules to attempt to meet or exceed adjacent channel or band interface requirements according to one or more communication protocols or standards. In an embodiment, the channel configuration  70 A may be used for a CDMA band five (V) signals where the TX band  73 A extends from 824 to 849 MHz ( 72 A,  72 B) and the RX band  73 B extends from 869 to 894 MHz ( 72 C,  72 D). In this configuration, The TX band  73 A and RX band  73 B are 25 MHz in width and separated by 20 MHz ( 72 C minus  72 B). As shown in  FIGS.  1 C to  1 G , the TX band  73 A may include a plurality of sub-bands  74 A,  74 B,  74 C and the RX band  73 B may including a plurality of sub-bands  75 A,  75 B,  75 C. In an embodiment, the sub-bands may be about 1.5 MHz wide (CDMA) and 5 MHz wide (W-CDMA). 
     In order to limit interface between adjacent bands, a filter module having a frequency characteristic  76 A as shown in  FIG.  1 D  may be applied to the TX band  73 A. Similarly, a filter module having a frequency characteristic  76 B as shown in  FIG.  1 E  may be applied to the RX band  73 B. As shown in  FIGS.  1 D and  1 E  the filter characteristics  76 A,  76 B ideally have a large dB rollout on either side of the communicated band (pass-band). The capacitors, inductors, and resistors required for such filter characteristics may be large and consume significant real estate when constructed on a dielectric wafer as known to those of skill in the art. One or more resonators may be employed to attempt to achieve a TX or RX signal 42 filter characteristic  76 A,  76 B. 
     Resonators may include surface acoustic wave (SAW) and bulk acoustic wave (BAW) devices. Such devices may be used in filters, oscillators and transformers and commonly cause the transduction of acoustic waves. In SAW and BAW, electrical energy is transduced to mechanical energy and back to electrical energy via piezoelectric materials. The piezoelectric materials may include quartz, lithium niobate, lithium tantalate, and lanthanum gallium silicate. One or more transverse fingers of conductive elements may be placed in the piezoelectric materials to convert electrical energy to mechanical energy and back to electrical energy. The SAW or resonator may include one or more one or more interdigital transducers (IDTs) (transverse fingers of electrical conductive elements) for such energy conversions or transductions. A resonator construction and material requirements may be more complex and expensive for electrical signals having high frequency content such as signals transmitted according to one or more RF communication protocols or standards. 
     It may be desirable for a filter or duplexer module  20  to generate frequency characteristics  76 C,  76 D specific to one or more sub-unit or bands of a TX or RX band  73 A,  73 B such as shown in  FIGS.  1 F,  1 G . Such duplexer modules  20  or filter modules may significantly suppress interface between TX and RX bands  73 A,  73 B and may be required for some communication protocols. In order to filter one or more sub-units  74 A,  74 B,  74 C,  75 A,  75 B,  75 C of a band  73 A,  73 B or different bands selectively (such as band I to V in a CDMA system), separate filters modules or duplexers may be required. 
       FIG.  2 A  is a block diagram of an electrical signal filter module  90 A including resonators according to various embodiments. The module  90 A includes three resonators  80 A,  80 B, and  80 C, resistors  94 A,  94 B, and a signal generator  92 A. In an embodiment, the signal generator  92 A may represent a TX signal to be communicated via an antenna  50 , the resistor  94 A may represent the load of the TX signal, and the resistor  94 B may represent the load of an antenna  50 . In an embodiment, the resonators  80 A,  80 B,  80 C form a T-shape between the signal to be transmitted and the antenna (source load  94 A and antenna load  94 B). The resonators  80 A,  80 B,  80 C may be SAW devices. A resonator  80 A,  80 B,  80 C commonly has a fixed resonate frequency and anti-resonate frequency similar to a pass band and stop band of a common inductor-capacitor type filter. 
     An acoustic wave resonator  80 A,  80 B,  80 C may be represented by corresponding electrical components according to various embodiments such as shown in  FIG.  2 B . As shown in  FIG.  2 B , a resonator  80 A may be represented by a first capacitor  82 A in parallel with a series coupling of an inductor  86 A, a second capacitor  82 B, and a resistor  84 A where the capacitors  82 A,  82 B may have a capacitance of Co, Cm, respectively, inductor  86 A may have an inductance of Lm and the resistor  84 A may have a resistance of Rm in an embodiment. Modeling of resonators or SAW devices via electrical components is described in the reference entitled “Surface Acoustic Wave Devices in Telecommunications: Modeling and Simulation” by Ken-Ya Hashimoto, published by Springer on Jul. 31, 2000, ISBN-10: 354067232X and ISBN-13: 978-3540672326. 
     The Cm and Lm may be related to the elasticity and inertia of an AW device  80 A,  80 B,  80 C. Co may represent the effective capacitance of the transverse electric fingers in the piezoelectric material of the AW  80 A,  80 B,  80 C. Rm may represent the heat generated by mechanical motion in the AW  80 A,  80 B,  80 C (the effective quality or Q limiter of the AW). Using the values Co, Cm, Lm, and Rm for first capacitor  82 A, inductor  86 A, second capacitor  82 B, and resistor  84 A, the resonance w r  and the anti-resonance w a  of an acoustic wave (AW) device  80 A may be defined by the following equations: 
     
       
         
           
             
               w 
               r 
             
             = 
             
               1 
               
                 
                   
                     
                       L 
                       m 
                     
                     
                       C 
                       m 
                     
                   
                 
               
             
             and 
               
             
               w 
               a 
             
             ≡ 
             
               1 
               
                 
                   
                     
                       
                         
                           L 
                           m 
                         
                         
                           C 
                           m 
                         
                         
                           C 
                           o 
                         
                       
                       
                         
                           
                             
                               C 
                               m 
                             
                             + 
                             
                               C 
                               o 
                             
                           
                         
                       
                     
                   
                 
               
             
             . 
           
         
       
     
     Using these equations AW  80 C may form a short path and the resultant filter formed by the AW  80 A, AW  80 B, and AW  80 C may have a pass band about the w r  of  80 A,  80 B and w a  of  80 C ( 77 C as shown in  FIG.  1 D ), a first notch before the pass band at w r  of  80 C ( 77 A in  FIG.  1 D ), and a second notch after the pass band at w a  of  80 A,  80 B ( 77 B in  FIG.  1 D ). These resonators AW  80 A,  80 B,  80 C resonate and anti-resonate values w r  and w a  are fixed as a function of the physical characteristics of the AW  80 A,  80 B,  80 C. 
     It may be desirable to shift the w r  and w a  of AW  80 A,  80 B,  80 C to shift the pass-band or stop-bands to tune to specific sub-bands  74 A,  74 B,  74 C,  75 A,  75 B,  75 C or different TX or RX bands  73 A,  73 B. It is also noted that the w r  and w a  of AW  80 A,  80 B,  80 C may vary as a function of the temperature of the AW, respectively. In such an embodiment it may be desired to correct for temperature variations accordingly. It is also noted that the w r  and w a  of AW  80 A,  80 B,  80 C may vary due to manufacturing variances, respectively. In such an embodiment it may be desirable to correct for manufacturing variances accordingly. In an embodiment various capacitors  98 A may be coupled in parallel or serially with a AW  80 A,  80 B,  80 C to be able to shift, tune, or modulate the w r  or w a  of the AW  80 A,  80 B,  80 C and accordingly its pass-band and stop-band(s). 
       FIGS.  2 C and  2 I  are block diagrams of modulated or tunable resonator modules  96 A,  96 G according to various embodiments. The module  96 A shown in  FIG.  2 C  may include a variable capacitor  98 A in parallel with an AW  80 A. Based on the above equations, the anti-resonate w a  may be modulated by the variable capacitor  98 A having a capacitance C v  (effective Co of an AW may be Co + C v  for module  96 A). The module  96 G shown in  FIG.  2 I  may include a variable capacitor  98 G in parallel with an AW  80 G and a variable capacitor  98 H in series with the AW  80 G. Based on the above equations, the anti-resonate w a  may be modulated by the variable capacitor  98 G having a capacitance C v1  and the variable capacitor  98 H having a capacitance C v2 . Similarly, the resonate w r  may be modulated by the variable capacitor  98 H having the capacitance C v2 . 
       FIG.  2 D  is a block diagram of an electrical signal filter module  90 B including tunable or modulated resonator modules  96 A,  96 B,  96 C according to various embodiments. The module  90 B is similar to module  90 A shown in  FIG.  2 A  in that it includes three resonators  80 A,  80 B, and  80 C in a similar T-configuration where the resonators  80 A,  80 B,  80 C have a fixed resonate frequency and anti-resonate frequency similar to a pass band and stop band of a common inductor-capacitor type filter where the anti-resonate frequency for each resonator  80 A,  80 B,  80 C is modulated or tuned by a variable capacitor  98 A,  98 B,  98 C. 
     As noted above AW  80 C may form a short path and the resultant filter formed by the AW  80 A, AW  80 B, and AW  80 C may have a pass band about the w r  of  80 A,  80 B and w a  of  80 C ( 77 C as shown in  FIG.  1 D ), a first notch before the pass band at w r  of  80 C ( 77 A in  FIG.  1 D ), and a second notch after the pass band at w a  of  80 A,  80 B ( 77 B in  FIG.  1 D ). By varying the capacitors  98 A,  98 B,  98 C, the pass band  77 C and second notch  77 B shown in  FIG.  1 D  may be varied. 
       FIGS.  2 E-H  are block diagrams of tunable filter modules including tunable or modulated resonators or AW that may be employed for various operations including filtering an RX band  73 B or sub-band  75 A,  75 B,  75 C in an embodiment. As shown in  FIG.  2 E  the tunable filter module  90 C may include tunable resonate or AW modules  96 D,  96 E,  96 F, and  96 G, resistor  94 C, and resistor  94 B. Similar to above, resistor  94 B may represent the antenna  50  load and resistor  94 C may represent a signal (RX or TX) load. In an embodiment, the module  90 C may include two tunable shorts  96 G and  96 F and two tunable pass AW modules  96 D,  96 E in series. Module  90 C is similar to module  90 A (T - configuration) with the addition of a second short  96 G that includes a capacitor  98 G designed to effect the anti-resonate frequency and a second tunable capacitor  98 H in series with the AW  80 G to further effect the resonate frequency of the AW  80 G. 
       FIG.  2 F  is a block diagram of an electrical signal filter module  90 D including a first tunable filter module  95 A and a second tunable filter module  95 B according to various embodiments. The module  90 D includes a first filter module  95 B, a second filter module  95 B, a first signal source  92 A and a resistor load  94 A, a second signal source  92 B and resistor load  94 C, and antenna load resistor  94 B. Module  95 A is similar to module  90 B and module  95 B is similar to module  90 C where module  95 A is a T-configuration module and module  95 B is a modified T-configuration with a second short (with a series tunable capacitor  98 H). In an embodiment, the module  90 D may be employed as a tunable duplexer  20  in  FIG.  1 A . 
       FIGS.  2 G,  2 H,  2 J  are block diagrams of tunable filter modules  95 C,  95 D,  95 E including tunable or modulated resonators or AW that may be employed for various operations including filtering a RX band  73 B or sub-band  75 A,  75 B,  75 C in an embodiment. As shown in  FIG.  2 G  the tunable filter module  90 E may include tunable resonate or AW modules  96 D,  96 G,  96 F, resistor  94 C, resistor  94 B, and effective capacitance  97 A,  97 B. Similar to above, resistor  94 B may represent the antenna  50  load and resistor  94 C may represent a signal (RX or TX) load and  92 B a signal source. In an embodiment, the module  90 E may include two shorts  96 G and  96 F and a single tunable AW module  96 D in series with the loads  94 C,  94 B. Module  90 E is similar to module  90 D with the elimination of the second module  96 E in series with the first module  96 D. 
     As shown in  FIG.  2 H  the tunable filter module  90 F may include tunable resonate or AW modules  96 D,  96 G,  96 F,  96 H, tunable capacitor  98 H, resistor  94 C, resistor  94 B, and effective capacitance  97 A,  97 B. Similar to above, resistor  94 B may represent the antenna  50  load and resistor  94 C may represent a signal (RX or TX) load and  92 B a signal source. In an embodiment, the module  90 F may include three tunable shorts  96 G,  96 F, and  96 H and a single tunable AW module  96 D in series with the loads  94 C,  94 B. Module  90 F is similar to module  90 F with the addition of a third short module  96 H. 
     As shown in  FIG.  2 J  the tunable filter module  95 E may include tunable resonate AW modules  96 B,  96 F, and a plurality of AW modules  80 A,  80 C,  80 D,  80 E,  80 G,  80 H,  80 I. In the filter module  95 E, tunable resonate AW modules  96 B,  96 F, and a plurality of AW modules  80 A,  80 C,  80 D,  80 E,  80 G,  80 H,  80 I form a series of “T” sub-filters such as  80 A,  96 B, and  80 C. As explained above each T sub-filter may create a frequency response with two passband (AW  80 A,  96 B) and a stopband ( 80 C). In the embodiment one or more AW  80 A to  80 I may not be tunable (AW modules  80 A,  80 C,  80 D,  80 E,  80 G,  80 H,  80 I in  FIG.  2 J ) while one or more AW  80 A to  80 I may be tunable ( 80 B and  80 F in  FIG.  2 J ). A tunable capacitor  98 B,  98 F may be coupled (in parallel) to an AW  80 A to  80 I when one or more AW  80 A to  80 I may be desirably tunable to modulate the AW  80 A to  80 I for temperature or process variations or provide frequency adjustments to the AW  80 A to  80 I. 
       FIGS.  3 A- 3 C  are diagrams of capacitor modules according to various embodiments where the modules may be used as capacitors  98 A to  98 G (in parallel to an AW  80 A to  80 F) and  98 H (in series with an AW  80 G). As shown in  FIG.  3 C , the module  120 C includes a single capacitor  104 A. The capacitor  104 A capacitance may be determined after the physical characteristics of an AW  80 A to  80 G are measured (to account for process variations or operating temperature variance). The capacitor  104 A capacitance may also be varied for different TX or RX bands  73 A,  73 B to be filtered by the module  96 A to  96 G including the module  120 C. 
     As shown in  FIG.  3 B , the module  120 B includes the capacitor  104 A and a second capacitor  104 B and resistor  106 A parallel to the first capacitor  104 A. The additional capacitor  104 B may further shift the AW  80 A to  80 G anti-resonate or resonate frequency to tune to a second band or sub-band. As shown in  FIG.  3 A , the module  120 A includes the capacitor  104 A, the second capacitor  104 B and a resistor  106 A parallel to the first capacitor  104 A, and a third capacitor  104 C and a second resistor  106 B parallel to the first capacitor  104 A (and second capacitor  104 B and resistor  106 A). The additional capacitor  104 C may still further shift the AW  80 A to  80 G anti-resonate or resonate frequency to tune to a third band or sub-band when the modules  120 A to  120 D are employed in parallel or series with a AW  80 A to  80 G as shown in modules  96 A to  96 G. 
       FIG.  3 D  is a diagram of a tunable capacitance module according to various embodiments. As shown in  FIG.  3 D , the module  120 D includes the capacitor  104 A, the second capacitor  104 B and resistor  106 A selectively parallel (via a switch  105 A) to the first capacitor  104 A, and a third capacitor  104 C and a second resistor  106 B selectively parallel (via the second switch  105 B) to the first capacitor  104 A (and second capacitor  104 B and resistor  106 A). The module  120 D may shift the AW  80 A to  80 H anti-resonate or resonate frequency to tune to a first, second, or third band or sub-band as a function of the switches  105 B,  105 A when coupled in parallel or series with the AW  80 A to  80 H as shown in modules  96 A to  96 G. The module  120 D may also shift an AW  80 A to  80 H anti-resonate or resonate frequency to account for temperature or manufacturing variants. 
       FIG.  3 E  is a diagram of a tunable capacitor module  600  according to various embodiments. The tunable capacitor module  600  includes a plurality of capacitor banks  602 , each switchable in operation via control lines  640 ,  642 , and  644 . In an embodiment each successive capacitor bank has twice the capacitance of the previous bank  602  so that each control line  640 ,  642 , and  644  is a digit of a binary number. In an embodiment, the capacitor banks are formed of CMOS FETs having their source and drain coupled via a resistor R DS  to effectively form capacitors in parallel. Each gate of the CMOS FETs  606 ,  608 ,  610 ,  612 ,  614  is coupled to the respective control lines  640 ,  642 ,  644 . Accordingly a tunable AW module  96 A to  96 G using the tunable capacitor  600  (in series or parallel) may have N 2  - 1 (where N is the number of control lines) different tunable anti-resonance or resonate frequencies based on the N 2  - 1 effective capacitances of the module  600 . Further details of digitally tunable capacitors are recited in commonly assigned PCT application entitled “METHOD AND APPARATUS FOR USE IN DIGITALLY TUNING A CAPACITOR IN AN INTEGRATED CIRCUIT DEVICE”, Attorney Docket - PER-024, Filed Mar. 2, 2009, and International Application Number PCT/US2009/001358, the entire contents of which are hereby incorporated herein by reference. 
       FIG.  4    is a block diagram of a configuration of a tunable filter module  130  including tunable resonators according to various embodiments. The filter module  130  may have a common circuit board or module  132 , a resonance or AW board or module  150 , and electrical component board or module  140 . The AW module  150  may include two or more resonators or AW  80 A,  80 B,  80 C,  80 I. In an embodiment, the AW  80 A,  80 B,  80 C may form the T-configuration  90 A shown in  FIG.  2 A . The AW module  150  may further include a bias AW  80 I. 
     The electrical component board or module  140  may include three tunable capacitors  98 A,  98 B,  98 C, a control logic module  146 , and an oscillator  144 . Each tunable capacitor  98 A,  98 B,  98 C may be coupled in parallel to an AW  80 A,  80 B,  80 C, respectively via two conductance lines  134  between the modules  140 ,  150 . Accordingly, the combination of an AW  80 A and a tunable capacitor  98 A may form a tunable AW module  96 A as shown in  FIG.  2 B . The oscillator  144  may be coupled to the bias AW  80 I via a conductance line  134 . The effective resonate frequency of the bias AW  80 I may modulate the oscillation of the oscillator  144  in a known and measurable way. 
     The control logic module  146  may receive control signals SPI for controlling the capacitance of tunable capacitors  98 A,  98 B, and  98 C and a stable clock or reference frequency (such a phase lock loop signal). In an embodiment, the AW  80 I resonate or anti-resonate frequencies may vary as function of temperature. Similarly the oscillator  144  frequency may vary as the AW  80 I resonate or anti-resonate frequencies fluctuate with temperature. The control logic  146  may monitor the change of oscillator frequency  144  via the stable reference frequency signal. The control logic  146  may then modulate the tunable capacitor’s capacitance based on known deltas to account for the oscillator frequency and thereby corresponding AW  80 A,  80 B,  80 C resonate or anti-resonate frequencies. In an embodiment, the delta may be added to the SPI control signals as needed to adjust for temperature effects of the AW  80 A,  80 B,  80 C. 
       FIG.  5 A  is a block diagram of an electrical signal filter module  190 A including switchable resonator modules (SRM) according to various embodiments. The module  190 A includes three switchable resonators modules (SRM)  180 A,  180 B, and  180 C, resistors  94 A,  94 B, and a signal generator  92 A. In an embodiment, the signal generator  92 A may represent a TX signal to be communicated via an antenna  50 , the resistor  94 A may represent the load of the TX signal, and the resistor  94 B may represent the load of an antenna  50 . In an embodiment, the switchable resonators modules (SRM)  180 A,  180 B,  180 C may form a T-shape between the signal to be transmitted and the antenna (source load  94 A and antenna load  94 B). The switchable resonators modules (SRM)  180 A,  180 B,  180 C may include one or more resonator devices or modules where one or more of the modules may include switchable resonators. The one or more resonators may have a fixed resonate frequency and anti-resonate frequency similar to a pass band and stop band of a common inductor-capacitor type filter. 
       FIGS.  5 B to  5 D  are block diagrams of SRM  184 A to  184 C according to various embodiments. As shown in  FIGS.  5 B to  5 D , a resonator module  184 A,  184 B,  184 C may include several (acoustic wave) resonators  82 A to  82 N where the resonators  82 A to  82 N may be bypassed or activated via one or more switches  182 A to  182 N. 
     In  FIG.  5 B  a switchable resonator module (SRM)  184 A may include two resonators  82 A,  82 B, and two switches  182 A,  182 B. The resonators  82 A,  82 B are coupled in series. A switch  182 A,  182 B may be coupled in parallel to resonator  82 A,  82 B, respectively. When a switch  182 A,  182 B is closed, the corresponding resonator  82 A,  82 B may be bypassed and inoperative. When a switch  182 A,  182 B is open, the corresponding resonator  82 A,  82 B may be active. In an embodiment each switch  182 A,  182 B may be controlled by a control signal S1A, S1B. In an embodiment, resonator  82 A and  82 B may operate exclusively or in tandem as a function of the control signals S1A, S1B. In a further embodiment a single signal may control the switches  182 A,  182 B where in a first signal state switch  182 A is open and switch  182 B is closed and in a second signal state switch  182 A is closed and switch  182 B is open. 
     In  FIG.  5 C  the switchable resonator module (SRM)  184 B includes three resonators  82 A,  82 B,  82 C and three switches  182 A,  182 B, and  182 C. The resonators  82 A,  82 B,  82 C are coupled in series. A switch  182 A,  182 B,  182 C may be coupled in parallel to a resonator  82 A,  82 B,  82 C, respectively. When a switch  182 A,  182 B,  182 C is closed the corresponding resonator  82 A,  82 B,  82 C may be bypassed and inoperative. Conversely when a switch  182 A,  182 B,  182 C is open, the corresponding resonator  82 A,  82 B,  82 C may be active. Each switch  182 A,  182 B,  182 C may be controlled by an independent control signal S1A, S1B, S1C. In an embodiment, resonators  82 A,  82 B, and  82 C may operate exclusively or in various combinations as a function of the control signals S1A, S1B, S1C. 
     In  FIG.  5 D  the switchable resonator module (SRM)  184 C includes a plurality of resonators  82 A to  82 N and corresponding switches  182 A to  182 N. The resonators  82 A to  82 N may be coupled in series. A switch  182 A to  182 N may be coupled in parallel to each resonator  82 A to  82 N, respectively. When a switch  182 A to  182 N is closed the corresponding resonator  82 A to  82 N may be bypassed and inoperative. Similarly, when a switch  182 A to  182 N is open the corresponding resonator  82 A to  82 N may be active. Each switch  182 A to  182 N may be controlled by a control signal S1A to S1N. In an embodiment, the resonators  82 A,  82 B, and  82 C may operate exclusively or in various combinations as a function of the control signals S1A to S1N. 
       FIG.  5 E  is a block diagram of a modulated or tunable resonator module system  190 B according to various embodiments. The tunable resonator module system  190 B includes several tunable resonator modules  196 A,  196 B,  196 C, forming a T configuration similar to  FIG.  5 A . Each tunable resonator module  196 A,  196 B,  196 C may include a variable capacitor  98 A,  98 B,  98 C coupled in parallel with a SRM  184 D,  184 E,  184 F. In each tunable modulator  196 A,  196 B,  196 C, the variable capacitor  98 A,  98 B,  98 C may modulate the anti-resonant frequency w a  of corresponding active resonators  82 A to  82 N,  83 A to  83 N, and  84 A to  84 N based on the capacitor’s selected capacitance C v  (effective capacitance C e  of an AW device may be equal to Co + C v  for a module  196 A). In an embodiment, the variable capacitor  98 A,  98 B,  98 C may module the anti- resonate w a  for each resonator  82 A to  82 N,  83 A to  83 N, and  84 A to  84 N not bypassed by switches  182 A to  182 N,  183 A to  183 B, and  185 A to  185 N where the switches are controlled by switch control signals S1A to S1N, S2A to S2N, and S3A to S3N. 
     In an embodiment each resonator  82 A to  82 N,  83 A to  83 N, and  84 A to  84 N may have a different resonance in each respective SRM  184 D,  184 E, and  184 F. The different resonances of the SRM  184 D,  184 E, and  184 F may enable a system  190 B to tune to different channels (different resonance frequencies) as shown in  FIGS.  6 A to  6 F  for frequency responses  197 A to  197 F. In an embodiment, the variable capacitor  98 A and  98 B in parallel with the SRM  184 D,  184 E may only module or tune the anti- resonate w a  of the active resonators  82 A to  82 N,  83 A to  83 N respectively. By selectively bypassing resonators  82 A to  82 N and  83 A to  83 N in the SRM  184 D,  184 E, the resonate frequency or effective pass-bands of the system  190 B may be tuned in addition to the stop bands. 
     In an embodiment control signals SxN in each corresponding SRM  184 D,  184 E,  184 F may be similarly opened or closed, e.g., control signals  182 A,  183 A, and  185 A may be simultaneously opened or closed (coordinated between modules  184 D,  184 E,  184 F). In a further embodiment the only one switch  182 A to  182 N,  183 A to  183 N,  185 A, to  185 N may be open at any time so only one resonator  82 A to  82 N,  83 A to  83 N,  84 A, to  84 N is active at any time. In an embodiment, the variable capacitor  98 C in parallel with the SRM  184 F may only module or tune the anti-resonate w a  of the active resonators  82 A to  82 N,  83 A to  83 N respectively. By selectively bypassing resonators  84 A to  84 N, the anti-resonate frequency or effective pass-bands of the SRM  196 C may be tuned in addition to the stop bands. 
       FIG.  5 F  is similar to  FIG.  5 E  except the tunable module  196 C is replaced by the module  96 G described with respect to  FIGS.  2 E and  2 I . The module  96 G may include a variable capacitor  98 G in parallel with an AW  80 G and a variable capacitor  98 H in series with the AW  80 G. Accordingly, the anti-resonate w a  of  96 G may be modulated by the variable capacitor  98 G having a capacitance C v1  and the variable capacitor  98 H having a capacitance C v2 . Similarly, the resonate w r  may be modulated by the variable capacitor  98 H having the capacitance C v2 . Capacitor  98 H may be subject to high voltages. 
       FIG.  5 G  is a block diagram of a modulated filter system  190 C similar to  FIG.  2 D  where the tunable resonators  96 A,  96 B,  96 C may be further tuned by series coupled variable capacitors  98 I,  98 J,  98 H. The variable capacitors  98 I and  98 J may modulate or tune the resonate frequencies of the resonators  80 A,  80 B, respectively. Such modulation may enable the system  190 C to tune different pass-bands and stop-bands as a function of the tunable capacitors  98 A,  98 B,  98 C,  98 I,  98 J, and  98 H. The tunable capacitors  98 I,  98 J in series with the resonators  80 A,  80 B may be subject to significant voltages, requiring the capacitors to be large. It is noted any resonator  80 A to  80 H shown in  FIGS.  2 A to  2 H  may be replaced by a SRM  184 A,  184 B, or  184 C such as shown in  FIGS.  5 B to  5 D . 
     In an embodiment it may be desirable to increase the isolation and stop-band rejection of a filter module.  FIG.  7 A  is a block diagram of a filter module  202 A according to various embodiments. The filter module  202 A includes an inductor  204 A and capacitor  206 A in series coupled in parallel to another inductor  204 B and capacitor  206 B in series. The inductors  204 A,  204 B may have an inductance L 1 , L 2  and the capacitors  206 A,  206 B may have a capacitance C 1 , C 2 . The filter module  202 A may have two pass bands at w 1  and w 2  surrounding a rejection point at w t . The rejection point may be limited by the quality, Q of the filter module  202 A. In the filter module  202 A the pass bands may be determined by the equations:  
     
       
         
           
             
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     As noted with reference to  FIG.  2 B , an AW  80 A may include an inductor  86 A in series with a capacitor  82 B with an inductance Lm and capacitance Cm, respectively. The resistor  84 A and capacitor  82 A may be nominal as a function of the inductor  86 A and capacitor  82 B. Accordingly, In an embodiment, the filter module  202 A may be represented by the parallel coupling of an AW  214 A,  214 B (the filter module  212 A shown  FIG.  7 B ). In this embodiment the acoustic wave module  214 A may represent the inductor  204 A and capacitor  206 A and the AW module  214 B may represent the inductor  204 B and capacitor  206 B of filter module  202 A. 
     The elasticity and inertia of an AW  214 A,  214 B may be configured or selected to have an equivalent Lm about L 1  or L 2  and Cm about C 1  and C 2  in an embodiment. In AW  214 A,  214 B, the parallel capacitance Co may represent the effective capacitance of the transverse electric fingers in the piezoelectric material and the resistance Rm may represent the heat generated by mechanical motion in the AW  214 A,  214 B (the effective quality or Q limiter of the AW). As a function of the signals to be filtered the pass bands and effective stop band between the pass bands w 1  and w 2  may need to be shifted or changed. 
     In an embodiment two or more inductor-capacitor filter modules (LCF)  202 A,  202 B, in series with a low resistive switch  205 A,  205 B may be coupled in parallel as shown  FIG.  8 A , filter module  208 A. The switches  205 A,  205 B may include one or more CMOS or MOSFET devices that have a low resistance when closed (as a function of a control signal S1A, S1B). In an embodiment, the LCF  202 A may have a first desired pass-band and stop-band and the LCF  202 B may have a second desired pass-band and stop-band. Via the control signals S1A, S2A a signal may be processed by either the LCF  202 A or the LCF  202 B of the filter module  208 A. Because the modules  202 A,  202 B are placed in parallel the operative signal path will only include the resistance of a single switch  205 A,  205 B, thus increasing the quality of the filter module  202 A,  202 B and its effective rejection strength (of its stop-band). 
     In an embodiment it may be desirable to process signals with larger voltage or limit circuit elements. The LCF  202 A,  202 B of filter module  208 A may be replaced by acoustic wave filters (AWF)  212 A,  212 B as shown in  FIG.  8 B , filter module  222 . Each AWF  212 A,  212 B may include two or more AW modules  214 A,  214 B coupled in parallel as shown in  FIG.  7 B . As noted and shown in  FIG.  8 C  a variable capacitor  218 A may be coupled in parallel in with AW device(s) or module(s) to provide adjustments for process variations in the AW device(s) or module(s) variations due to temperature, and enable shifting of pass-band or stop-bands of the device(s). As shown in the filter module  224  of  FIG.  8 C , a variable capacitor  218 A may also be placed in parallel with one or more AWF  212 A,  212 B. In filter module  224 , the capacitor  218 A capacitance may be varied as a function of the switch  216 A,  216 B control signals S1A, S1B to modulate the AWF  212 A or the AWF  212 B. 
       FIG.  9 B  is a block diagram of filter module  230 A according to various embodiments. The filter module  230 A may include a first capacitive-tunable, parallel switched AW module filter  232 A, a second capacitive-tunable, parallel switched AW module filter  232 B, a first capacitive-tunable parallel switched AWF module filter  224 A, a capacitive-tunable AW module  234 A, and impedance inversion modules  228 A,  228 B. The module  232 A may be coupled to the module  232 B via the inversion module  228 A and the module  232 B may be coupled to the module  224 A via the inversion module  228 B. The module  234 A may be coupled to ground and the module  232 A. 
     In an embodiment, the first capacitive-tunable, parallel switched AW module filter  232 A may include AW modules  214 A,  214 B, switches  216 A,  216 B, and variable capacitor  218 A. AW module  214 A is series coupled to switch  216 A and AW module  214 B is series coupled to switch  216 B. Each module, switch pair  214 A,  216 A,  214 B,  216 B is coupled in parallel to the variable capacitor  218 A. Similarly, the second capacitive-tunable, parallel switched AW module filter  232 B may include AW modules  214 C,  214 D, switches  216 C,  216 D, and a variable capacitor  218 B. AW module  214 C is series coupled to switch  216 C and AW module  214 D is series coupled to switch  216 D. Each module, switch pair  214 C,  216 C,  214 D,  216 D is coupled in parallel to the variable capacitor  218 B. 
     The capacitive-tunable, parallel switched AWF module filter  224 A may include AWF modules  212 A,  212 B, switches  216 E,  216 F, and variable capacitor  218 C. AWF module  212 A is series coupled to switch  216 E and AWF module  212 B is series coupled to switch  216 F. Each module, switch pair  212 A,  216 E,  212 B,  216 F is coupled in parallel to the variable capacitor  218 C. Each AWF module  212 A,  212 B includes two parallel coupled AW modules  214 C,  214 D and  214 E,  214 F, respectively. The capacitive-tunable AW module  234 A includes an AW module  214 G coupled in parallel to a variable capacitor  218 D. 
     In an embodiment, the inversion module  228 A,  228 B may be a K-filter  228  as shown in  FIG.  9 A . The filter  228  includes two capacitors  226 A,  226 B in series with a third capacitor  226 C in parallel and between the series pair  226 A,  226 B. In an embodiment, the capacitors  226 A,  226 B have a capacitance of -C and the capacitor  226 C has a capacitance of +C. As shown in the  FIG.  9 B , the capacitor  226 C of the inversion modules  228 A,  228 B is also coupled to ground. 
     In an embodiment, the module  234 A may provide a fixed high rejection and tunable pass-band, the modules  232 A,  232 B may provide a movable, switchable pass-band and tunable rejection band, and the module  224 A may provide a movable, switchable high rejection point and pass-band. The filter module  230 A of  FIG.  9 B  may be employed to generate the frequency responses  240 A,  240 B shown in  FIGS.  10 A,  10 B  where the control signals S1A, S1C, S1E may be active, inactive while the control signals S1B, S1D, S1F may be inactive, active, respectively to shift the pass-bands and stop or rejection bands shown in  FIGS.  10 A,  10 B  ( 240 A,  240 B). In an embodiment  230 B shown in  FIG.  9 C  the inversion modules  228 A,  228 B of  FIG.  9 B  may be replaced by one or more capacitors  226 D,  226 E coupled to ground. 
       FIG.  11    is a diagram of filter frequency responses  250  according to various embodiments.  FIG.  11    depicts a first frequency response  258 B and a second frequency response  258 A. In an embodiment a filter response  258 A,  258 B includes a passband  261  with a passband edge  262  and stopband  263 . Further a filter response  258 A,  258 B may have a maximum acceptable loss  252  in the passband area  261  (creating the passband edge  262 ) and a minimum attenuation or rejection  256  in the stopband  263 . Further the minimum attenuation or rejection  256  in the stopband  263  may need to be achieved by a particular frequency  254  such a channel boundary or cutoff frequency. In an embodiment a filter mechanism or module such as resonator module  292 B of  FIG.  13 B  may produce a first frequency response  258 B during ideal operation and fabrication conditions. The same filter module  292 B may generate the shifted frequency response  258 A due to non-ideal operation or fabrication conditions. In an embodiment, the frequency response shift from  258 B to  258 A may be due to temperature fluctuations and fabrication variations. 
     Given the potential filter module  292 B frequency response shift (from  258 B to  258 A), the passband  261  region or width of a signal processed by the filter module  292 B may be narrowed or reduced to ensure that the minimum required attenuation  256  is achieved by a required frequency  254 . The required frequency  254  may be the start of another channel and the filter module  292 B may be required to prevent signal leakage into adjacent channels. The distance between the channel boundary  254  and passband edge  262  is commonly termed the guard band of a filter or channel. In a system or architecture such as channel architecture  310 A,  310 B,  310 C shown in  FIGS.  15 A,  15 B,  15 C  the guard band ( 316 B in  FIGS.  15 B and  318 B  in  FIG.  15 C ) represents lost or unusable bandwidth. Accordingly it may be desirable to minimize the guard band  316 B,  318 B by reducing the effect of temperature and process or fabrication variations of filters or filter architectures that may be employed to limit or prevent signal leakage between adjacent channels ( 312 A,  314 A, and  312 B). 
       FIG.  12    is a flow diagram of a filter configuration method  270  according to various embodiments. In the method  270  the maximum passband loss  252  may be selected where this loss level may be required or indicated (by a standard or other communication protocol establishment organization) (activity  272 ). The filter response stopband minimum attenuation  256  needed to reduce or limit signal leakage into adjacent channels may be selected where the minimum attenuation may be required or indicated (by a standard or other communication protocol establishment organization) (activity  274 ). Further the minimum stopband edge  254  for the minimum attenuation  256  may also be selected where the minimum stopband edge  254  may be required or indicated (by a standard or other communication protocol establishment organization) (activity  276 ). 
     In the method  270  the minimum stopband edge  254  of a non-tunable filter  292 B may be pre-shifted to ensure the filter response  258 B when shifted due to temperature or process variations achieves the minimum attenuation  256  by the desired or required boundary or edge  254  (activity  278 ). Further, the filter passband  262  edge may also be shifted, effectively reducing the usable signal bandwidth to ensure less than the maximum loss  252  is present in the passband (activity  282 ). Accordingly the effective guard band  316 B,  318 B may be increased. 
       FIG.  13 A  is a simplified block diagram of a filtering architecture  290 A according to various embodiments. The filter architecture includes a filter  292 A coupled in series with a tunable filter  294 A. In an embodiment, the filter  292 A may have a desired frequency response shown as  258 A shown in  FIG.  11    but be subject to temperature or process variations where the fixed filter  292 A frequency response may shift to the filter response  300 A shown in  FIG.  14 A . Such a worst case frequency response  302 A may be unacceptable due to potential signal leakage beyond the desired channel or signal boundary  254 . The frequency response  302 A otherwise has stable passband and stopband  304 A. 
     The tunable filter  294 A may have a tunable frequency response such as module  294 B shown in  FIG.  13 B  where temperature and process variations are corrected or modulated by an adjustable element such as a tunable capacitor  218 A. The tunable filter  294 A may have a frequency response  300 B in  FIG.  14 B . As shown in  FIG.  14 B  the frequency response  302 B may achieve the desired or required maximum passband loss  252  with an edge  262  than is greater in frequency than the filter  302 A (when adjusted to account for potential shifts) and correspondingly a smaller needed guard band  316 B,  318 B. The filter response  302 B for tunable filter  294 A may also meet the minimum attenuation  256  by the frequency boundary  254  (point  303 B in  FIG.  14 B ). The tunable filter  294 A filter response  302 B may have a second, unacceptable passband  304 B within the adjacent channel  305  and thus be unacceptable as a single filter. 
     In an embodiment, the filter module  292 A,  292 B and tunable filter  294 A,  294 B,  290 A,  290 B respectively, in combination may create the frequency response  300 C shown in  FIG.  14 C . As shown in  FIG.  14 C  the net frequency response  300 C may include the desirable stopband of filter  294 A, B without the subsequent passband  304 B due the filter  292 A, B stopband  304 A. Further, while the filter  292 A, B stopband edge  303 A may vary with temperature and process variations it is sufficient to suppress the filter  294 A, B undesirable second passband  304 B. The resultant frequency response  300 C may have an acceptable passband loss  252  and minimum stopband attenuation  256  by the desired boundary or frequency cutoff  254  without temperature and process variations. 
       FIG.  13 B  is a block diagram of a filter architecture  290 B including a modulated or tunable resonator module  294 B and a resonator module  292 B according to various embodiments. The resonator module  292 B may be a non-tunable filter that may be configured to a frequency response similar to frequency response  300 A shown in  FIG.  14 A . The resonator module  292 B may include surface acoustic wave (SAW) and bulk acoustic wave (BAW) devices where the device enables the transduction of acoustic waves. In an acoustic wave device electrical energy is transduced to mechanical energy back to electrical energy via piezoelectric materials. The piezoelectric materials may include quartz, lithium niobate, lithium tantalate, and lanthanum gallium silicate. One or more transverse fingers of conductive elements may be placed in the piezoelectric materials to convert electrical energy to mechanical energy and back to electrical energy. 
     In an embodiment, the tunable resonator  294 B may include one or more acoustic wave modules or devices  214 A,  214 B, and a tunable capacitor  218 A. The AW modules  214 A,  214 B, and tunable capacitor  218 A may be coupled in parallel in an embodiment as shown in  FIG.  13 B . As noted this configuration may have two pass bands at w 1  and w 2  surrounding a rejection point at w t. . The pass bands at w 1  and w 2  may correspond to filter response components  302 B and  304 B shown in  FIG.  14 B  and the rejection point at w t  may correspond to the component  303 B. The variable capacitor  218 A coupled in parallel with the AW modules  214 A,  214 B may tune or modulate the filter module  294 B frequency response  300 B to correct for temperature or process variations. Other resonator filters such as shown in  FIGS.  2 A to  2 H ,  FIG.  4   ,  FIGS.  5 A to  5 G ,  FIGS.  7 B to  8 C , and  FIGS.  9 B- 9 C  may be employed in whole or part as a tunable resonator or filter  294 B. 
     The filter architecture  290 A may be modified such as shown in  FIGS.  16 A,  16 B,  16 C, and  16 D  for different filter requirements or parameters. As shown in  FIG.  16 A , the filter architecture  330 A may include a switchable and tunable filter module  334 A. Such a module  334 A and resulting architecture (and switchable frequency response) may be employed in communication architectures requiring varying filters to process one or more signals. As shown in  FIG.  16 B , a switchable, tunable filtering architecture  330 B may include a first switchable tunable filter module  335 A and a second switchable tunable filter module  335 B. Each module  335 A,  335 B may include a filter module  332 B,  332 C similar to module  292 B ( FIG.  13 B ). Each switchable, tunable module  335 A,  335 B may also include an AWF  212 A, AWF  212 B, switch pairs  216 E,  217 E and  216 F,  217 F, and a AW module  96 C,  96 F. 
     Each AWF module  212 A,  212 B may include two AW modules  214 C, 2124D, and  214 E,  214 F coupled in parallel and a variable capacitor  218 C,  218 D further coupled in parallel to the two AW modules  214 C,  214 D and  214 E,  214 F, respectively. The tunable modules  335 A,  335 B may include the AW module  96 C,  96 D located between the AW  332 B,  332 C and  212 A,  212 B and ground. Each AW module  96 C,  96 D may include an AW module  80 C,  80 F and a tunable capacitor  98 C,  98 F coupled in parallel to the AW module  80 C,  80 F. Each switchable, tunable module  335 A,  335 B may be coupled in parallel. As noted above each AWF module  212 A,  212 B may have a frequency response that includes two pass bands at w 1  and w 2  surrounding a rejection point at w t . In an embodiment, the switchable, tunable architecture  330 B may operate in two modes: mode 1 (switch pair  216 E,  217 E closed and switch pair  216 F,  217 F open) (frequency responses  320 A and  320 B shown in  FIGS.  17 A and  17 B  may combine to create response  320 C shown in  17 C) and mode 2 (switch pair  216 E,  217 E open and switch pair  216 F,  217 F closed), frequency response  320 D and  320 E shown in  FIGS.  17 D and  17 E  may combine to create response  320 F shown in  17 F. 
     The AW module  332 B,  332 C may have a frequency response  320 A,  320 D shown in  FIG.  17 A ,  FIG.  17 D , respectively. When this frequency response  320 A,  320 D is combined with the switchable, tunable AW module’s  335 A frequency response mode 1  320 B -  FIG.  17 B  or tunable AW module’s  335 B, mode 2  320 D -  FIG.  17 E , the resultant frequency response may be combined mode 1  320 C -  FIG.  17 C  or mode 2  320 F -  FIG.  17 F . Such a switchable, tunable filter architecture  330 A,  330 B may be applied in a channel architecture requiring different filter operation modes such as shown in  FIGS.  15 A to  15 C . The AWF  96 C,  96 F may provide an additional stop band as a function of the AW  80 C,  80 F configuration. 
     In the channel configuration  310 A shown in  FIG.  15 A  a time division multiplex (TDD) band 38 is located between a transmit channel of band 7 and a receive channel of band 7. In an embodiment band 7 may represent frequency division duplex (FDD) spectrum of a long term evolution (LTE) system and band 38 may represent TDD spectrum of the LTE system or architecture. In the combined LTE FDD, TDD spectrum, band 38 spectrum  314 A may be sandwiched between band 7’spectrum  312 A  312 B. When the TDD channel or band 38 is transmitting (as shown in configuration  310 B shown in  FIG.  15 B ) band 38 should not leak into RX band 7  312 B. In band 38 transmit mode  310 B, mode 1 of the filter architecture  330 B may be employed to generate the frequency response  320 C shown in  FIG.  17 C . 
     In channel configuration  310 B during band 38 transmit mode, a guard band  316 B may be located between band 38’s transmit section or passband  316 A and band 7’s receive band  312 B. In mode 1 the filter architecture  330 B may generate the frequency response  320 C shown in  FIG.  17 C  where the stopband  324 A is located in the guard band  316 B. When band 38 is in receive mode ( FIGS.  15 C,  310 C ), the band 7 transmit channel  312 A may interfere with the band 38 receive channel  318 A. In such a configuration the filter architecture  330 B of  FIG.  16 B  may operate in the second mode (mode 2) to generate the frequency response  320 F shown in  FIG.  17 F . The frequency response  320 F stopband  324 B may be located in the guard band  318 B when band 38 is in receive mode. The architecture  330 B shown in  FIG.  16 B  may reduce the guard band size  316 B,  318 B enabling greater bandwidth utilization (of band 38 in the embodiment shown in  FIGS.  15 A to  15 C ). 
     Another filter embodiment  330 C is shown in  FIG.  16 C . Filter  330 C includes a first, tunable switchable filter module  334 C and a second, tunable switchable filter module  334 D serially coupled. The first, tunable switchable filter module  334 C may include a first resonator  332 B, a first tunable resonator  212 A, a first, grounded tunable resonator  96 C, and a first opposite switch pair  216 E,  217 E. The switch  217 E, the first resonator  332 B, and the first tunable resonator  212 A may be serially coupled together and the serial group ( 217 E,  332 B,  212 A) may be coupled in parallel to the switch  216 E. The AWF module  96 C may be located between the AW  332 B and  212 A and ground. The AWF  96 C may include an AW module  80 C and a tunable capacitor  98 C coupled in parallel to the AW module  80 C. 
     Similarly, the second, tunable switchable filter module  334 D may include a second resonator  332 C, a second tunable resonator  212 B, a second, grounded tunable resonator module  96 F, and a second opposite switch pair  216 F,  217 F. The switch  217 F, the second resonator  332 C, and the second tunable resonator  212 B may be serially coupled together and the serial group ( 217 F,  332 C,  212 B) may be coupled in parallel to the switch  216 F. The AWF module  96 F may be located between the AW  332 C and  212 B and ground. The AWF  96 F may include an AW module  80 F and a tunable capacitor  98 F coupled in parallel to the AW module  80 F. 
     The filter module  334 C, when active (switch  216 E open, switch  217 E closed, switch  216 F closed, switch  217 F open (mode 1)) may produce the frequency response  320 C shown in  FIG.  17 C . The filter module  334 D, when active (switch  216 E closed, switch  217 E open, switch  216 F open, switch  217 F closed (mode 2)) may produce the frequency response  320 F shown in  FIG.  17 F . In another mode, mode 3 switches  216 E and  216 F may both be open and switches  217 E,  217 F closed (engaging both filter modules  334 C,  334 D) generating the frequency response  320 G shown in  FIG.  17 G . Such a frequency response may be employed to protect bands on either side of the combined filter, such as band 7 transmit  312 A and receive  312 B shown in  FIG.  15 A . The AWF  96 C may provide an additional stop band as a function of the AW  80 C configuration. 
     The filter system or architecture  330 C may have an unacceptable insertion loss in mode 1 or 2 given the potential loss and capacitance of the open switches  216 F,  217 E (mode 2), switch  216 E,  217 F (mode 1). Another filter architecture  330 D enabling modes 1, 2, and 3 with a lower insertion loss is show in  FIG.  16 D . As shown in  FIG.  16 D , the filter architecture  334 E includes a first filter module  336 A, a second filter module  336 B, and a third filter module  336 C, all coupled in parallel to each other. The first filter module  336 A includes a first resonator  332 B, a first AWF  212 A, a first, grounded AWF  96 C, and a switch pair  216 E,  217 E coupled in series where these resonators in series may produce the frequency response  320 C shown in  FIG.  17 C  (mode 1 - switch pair  216 E,  217 E closed, switch pair  216 F,  217 F open, and switch pair  216 G,  217 G open). 
     The second filter module  336 B includes a second resonator  332 C, a second AWF  212 B, a second, grounded AWF  96 F, and a switch pair  216 F,  217 F coupled in series where these resonators in series may produce the frequency response  320 F shown in  FIG.  17 F  (mode 2 -switch pair  216 E,  217 E open, switch pair  216 F,  217 F closed, and switch pair  216 G,  217 G open)). The third filter module  336 C may include the first resonator  332 B, the first AWF  212 A, the second resonator  332 C, the second AWF  212 B, the first, grounded AWF  96 C, the second, grounded AWF  96 F, and the switch pair  216 G,  217 G in series. In mode 3, the combined resonators  332 B,  212 A,  332 C, and  212 B may generate the frequency response  320 G shown in  FIG.  17 G . 
     A signal processing architecture  330 E is shown in  FIG.  16 E . The architecture  330 E may include a first filter system  215 A, a second filter system  215 B, a two position switch  216 H, a power amplifier (PA)  12 , a low noise amplifier (LNA)  14 , an antenna  50 , and a mixer  60 A. A signal  8  to be transmitted via antenna  50  may be amplified by PA  12  to produce an amplified signal  22 . The resultant amplified signal  22  may include signal content beyond the desired or permitted transmission bandwidth such as band 38 transmit channel  316 A shown in  FIG.  15 B . The resultant signal  22  may filtered by the filter system  215 A. The filter system  215 A may include the first resonator module  332 B, a first grounded resonator module  96 C (including a resonator  80 C and a tunable capacitor  98 C), and a first parallel resonator module (including resonator  214 C,  214 D and a tunable capacitor  218 C). In an embodiment, the first filter system  215 A may generate the frequency response  320 C shown in  FIG.  17 C . 
     The filtered, amplified signal may be coupled to the antenna  50  via the switch  216 H. Similarly a signal  42  received on the antenna  50  may be filtered by the second filter system  215 B. The filter system  215 B may include the second resonator module  332 C, a second grounded resonator module  96 F (including a resonator  80 F and a tunable capacitor  98 F) and a second parallel resonator module (including resonator  214 E,  214 F and a tunable capacitor  218 D). In an embodiment, the second filter system  215 B may generate the frequency response  320 F shown in  FIG.  17 F . The resultant filtered, received signal may be amplified by the LNA  14 . The amplified, filtered, received signal may be shifted to another center frequency (such as base-band) via the mixer  60 A and a reference frequency signal  60 B to generate the frequency shifted, amplified, filtered, received signal  60 C.The filter architecture  330 E may be employed in a TDD communication system such as band 38 in an LTE spectrum in an embodiment. 
     In an embodiment, the method  340  shown in  FIG.  18    may be employed to configure a filter architecture  290 A,  290 B, 330A-E shown in  FIGS.  13 A,  13 B, and  16 A- 16 E , respectively. In method  340  the maximum insertion loss (passband maximum loss)  252  may be selected (as required or indicated) (activity  342 ). The stopband minimum edge(s)  254  may then be selected (as required or indicated) (activity  344 ). Similarly the minimal attenuation for the stopband edge may also be selected (as required or indicated)  256  (activity  346 ). Based on these requirements  252 , 254 ,  256 , a tunable resonator filter  294 A,  294 B,  334 A,  334 B may be configured to have a stopband located at the point  254  and having at least the minimum attenuation  256  while meeting the maximum passband loss  252  requirement (activity  348 ). A resonator filter  292 A,  292 B,  332 A,  332 B,  332 C may be configured to have stopband extend pass the initial stopband  254  with the minimum attenuation  256  and the maximum passband loss  252  based on the potential temperature and process variation of the filter (activity  352 ). Activities  348 ,  352  may be performed in any order or contemporaneously. 
       FIG.  19 A  is a block diagram of an electrical signal filter module  360 A including resonators  80 A,  80 B,  80 C and diagrams of filter frequency responses  362 A,  362 B,  362 C of resonators  80 A,  80 B,  80 C, respectively according to various embodiments. A resonator  80 A,  80 B,  80 C may be represented by corresponding electrical components according to various embodiments such as shown in  FIGS.  2 B,  20 A,  20 B . As shown in  FIGS.  2 B,  20 A,  20 B , a resonator  80 A,  80 B,  80 C may be represented by a first capacitor  81 A,  81 B,  81 C in parallel with a series coupling of an inductor  86 A,  86 B,  86 C, second capacitor  82 A,  82 B,  82 C, and resistor  84 A,  84 B,  84 C where the capacitors  81 A,  81 B,  81 C,  82 A,  82 B,  82 C may have a capacitance of C OA , C OB , C OC , C MA , C MB , C MC , respectively, inductors  86 A,  86 B,  86 C may have an inductance of L MA , L MB , L MC  and the resistors  84 A,  84 B,  84 C may have a resistance of R MA , R MB , R MC  in an embodiment. 
     The values of C MA , C MB , C MC  and L MA , L MB , L MC  may be related to the elasticity and inertia of an AW  80 A,  80 B,  80 C in an embodiment. The values of C OA , C OB , C OC  may represent the effective capacitance of the transverse electric fingers in the piezoelectric material of the AW  80 A,  80 B,  80 C in an embodiment. The values of R MA , R MB , R MC  may represent the heat generated by mechanical motion in the AW  80 A,  80 B,  80 C (the effective quality or Q limiter of the AW) in an embodiment. Using the values C OA , C MA , L MA , and R MA  for the first capacitor  81 A, the inductor  86 A, the second capacitor  82 B, and the resistor  84 A for resonator  80 A, the resonance w r  and the anti-resonance w a  of the acoustic wave (AW) device  80 A may be defined by the following equations: 
     
       
         
           
             
               w 
               
                 r 
                 1 
               
             
             ≡ 
             
               1 
               
                 
                   
                     
                       L 
                       
                         M 
                         A 
                       
                     
                     
                       C 
                       
                         M 
                         A 
                       
                     
                   
                 
               
             
             and 
               
             
               w 
               
                 a 
                 1 
               
             
             ≡ 
           
         
       
     
     
       
         
           
             
               1 
               
                 
                   
                     L 
                     M 
                     A 
                     C 
                     M 
                     A 
                     C 
                     O 
                     A 
                     / 
                     C 
                     M 
                     A 
                     + 
                     C 
                     O 
                     A 
                   
                 
               
             
               
           
         
       
     
     Using these equations the AW  80 A may form the frequency response  362 A shown in  FIG.  19 A , the response similar to a low pass filter with a pass band about the resonate frequency, f r1  and stop band about the anti-resonance f a1 . Similarly, the AW  80 B may form the frequency response  362 B shown in  FIG.  19 A , the response similar to a low pass filter with a pass band about the resonate frequency, f r2  and stop band about the anti-resonance f a2 . The AW  80 C may form a short path and its frequency response  362 C shown in  FIG.  19 A  may be similar to a high pass filter with a pass band about the anti-resonance f a3  and stop band about the resonate frequency, f r3 . It is noted that the resonator AW  80 A,  80 B,  80 C resonate and anti-resonate frequencies f r1 , f r2 , f r3  and f a1 , f a2 , f a3  may be fixed as a function of the physical characteristics of the AW devices  80 A,  80 B,  80 C. Using the resultant frequency response of an AW device  80 A,  80 B,  80 C based on its physical characteristics, various filter responses may be formed by various combinations of the devices  80 A,  80 B,  80 C. 
       FIG.  19 B  is a diagram of filter frequency responses  362 A,  362 B,  362 C of the electrical signal filter module  360 A including resonators  80 A,  80 B,  80 C of  FIG.  19 A  in a first, pass-band filter configuration  364 A having a center frequency f c  according to various embodiments.  FIG.  19 D  is a diagram of the effective combination of filter frequency responses  362 A,  362 B,  362 C of the electrical signal filter module  360 A including resonators  80 A,  80 B,  80 C of  FIG.  19 A  in the first, pass-band filter configuration  364 C having a center frequency f c  according to various embodiments. 
     In  FIGS.  19 B and  19 D  the AW device  80 A frequency response  362 A resonate frequency, f r1  may be configured to be greater than f c  of the filter  364 A and accordingly its stop band about the anti-resonance f a1  also greater than f c  of the filter  364 A and its resonate frequency, f r1 . Similarly, the AW device  80 B frequency response  362 B resonate frequency, f r2  may be configured to be greater than f c  of the filter  364 A and the AW device  80 A frequency response  362 A resonate frequency, f r1 . The AW device  80 B stop band about its anti-resonance f a2  may also be greater than f c  of the filter  364 A, its resonate frequency, f r2  and the AW device  80 A resonate frequency, f r1  and anti-resonate frequency, f a1 . The short part AW device  80 C  frequency response  362 C anti-resonate frequency, f a3  may be configured to be less than f c  of the filter  364 A and accordingly its stop band about the resonance f r3  also less than f c  of the filter  364 A and its anti-resonate frequency, f a3 . As shown in  FIG.  19 D  the effective combination of the AW devices  80 A,  80 B,  80 C having the frequency responses  362 A,  362 B,  362 C as shown in  FIG.  19 B  (based on the AW devices physical characteristics) may form the band pass filter  364 C with bandwidth  366 A. 
       FIG.  19 C  is a diagram of filter frequency responses  362 A,  362 B,  362 C of the electrical signal filter module  360 A including resonators  80 A,  80 B,  80 C of  FIG.  19 A  in a notch filter configuration  364 B having a center frequency f c  according to various embodiments.  FIG.  19 E  is a diagram of the effective combination of filter frequency responses  362 A,  362 B,  362 C of the electrical signal filter module  360 A including resonators  80 A,  80 B,  80 C of  FIG.  19 A  in the notch filter configuration  364 E having a center frequency f c  according to various embodiments. 
     In  FIGS.  19 C and  19 E  the AW device  80 A frequency response  362 A anti-resonate stop-band frequency, f a1  may be configured to be less than f c  of the filter  364 A and accordingly its pass band about the resonance f r1  also less than f c  of the filter  364 A and its anti-resonate frequency, f a1 . The AW device  80 B frequency response  362 B anti-resonate frequency, f a2  may be configured to be about the center frequency, f c  of the filter  364 B and greater than the AW device  80 A frequency response  362 A anti-resonate frequency, f a1 . The AW device  80 B pass band about its resonance f r2  may also be less than f c  of the filter  364 B, its anti-resonate frequency, f a2  and the AW device  80 A anti-resonate frequency, f a1 . The AW device  80 B pass band about its resonance f r2  may be greater the AW device  80 A resonate frequency, frl. 
     The short part AW device  80 C frequency response  362 C stop-band resonate frequency, f r3  may be configured to be greater than f c  of the filter  364 A and accordingly its pass-band about the anti-resonance f a3  also greater than f c  of the filter  364 A and its resonate frequency, f r3 . As shown in  FIG.  19 E  the effective combination of the AW devices  80 A,  80 B,  80 C having the frequency responses  362 A,  362 B,  362 C as shown in  FIG.  19 C  (based on the AW devices physical characteristics) may form the notch filter  364 D with bandwidth  366 B. 
       FIG.  21 A  is a block diagram of a tunable electrical signal filter module  380 A including resonators  80 A,  80 C,  80 D, variable capacitors  98 A,  98 C, and  98 D, and diagrams of filter frequency responses  362 A,  362 C,  362 D of resonators  80 A,  80 C,  80 D, respectively according to various embodiments. In an embodiment, the variable capacitor  98 A may be coupled in parallel to the AW device  80 A. The variable capacitor  98 C may be coupled in series with the AW device  80 C. The variable capacitor  98 D may be coupled in series with the AW device  80 D. The AW device  80 C coupled in series with the variable capacitor  98 C may form a first short path. The AW device  80 D coupled in series with the variable capacitor  98 D may form a second short path. 
     Similar to  FIG.  19 A  the AW  80 A may form the frequency response  362 A shown in  FIG.  21 A , the response similar to a low pass filter with a pass band about the resonate frequency, f r1  and stop band about the anti-resonance f a1 . The AW  80 C may form a short path and its frequency response  362 C shown in  FIG.  21 A  may be similar to a high pass filter with a pass band about its anti-resonance f a3  and a stop band about its resonate frequency, f r3 . The AW  80 D may also form a short path and its frequency response  362 D shown in  FIG.  21 A  may be similar to a high pass filter with a pass band about its anti-resonance f a4  and a stop band about its resonate frequency, f r4 . 
     It is noted that the resonator AW devices  80 A,  80 C,  80 D resonate and anti-resonate frequencies f r1 , f r3 , f r4  and f a1 , f a3 , f a4  may be fixed as a function of the physical characteristics of the AW devices  80 A,  80 C,  80 D. The variable capacitors  98 A,  98 C,  98 D may shift the device  80 A,  80 C,  80 D characteristics as described above. Using the resultant frequency response of a AW device  80 A,  80 C,  80 D based its physical characteristics various filter responses may be formed by various combinations of the devices  80 A,  80 C,  80 D. 
       FIG.  21 B  is a diagram of filter frequency responses  362 A,  362 C,  362 D of the electrical signal filter module  380 A ( FIG.  21 A ) including resonators  80 A,  80 C,  80 D of  FIG.  21 A  in a notch filter configuration  380 B having a center frequency f c  according to various embodiments.  FIG.  21 C  is a diagram of the effective combination  380 C of filter frequency responses  362 A,  362 C,  362 D of the electrical signal filter module  380 A including resonators  80 A,  80 C,  80 D of  FIG.  21 A  in the notch configuration  380 C having a center frequency f c  and bandwidth  386 C according to various embodiments. 
     In  FIGS.  21 B and  21 C  the AW device  80 A frequency response  362 A anti-resonate stop-band frequency, f a1  may be configured to be less than f c  of the filter  380 B and accordingly its pass band about the resonance f r1  also less than f c  of the filter  380 B and its anti-resonate frequency, f a1 . The short part AW device  80 C frequency response  362 C stop-band resonate frequency, f r3  may be configured to be about the f c  of the filter  380 A and accordingly its pass-band about the anti-resonance f a3  greater than f c  of the filter  380 A and its resonate frequency, f r3 . The second short part AW device  80 D frequency response  362 D stop-band resonate frequency, f r4  may be configured to be greater than the f c  of the filter  380 A and accordingly its pass-band about the anti-resonance f a3  greater than f c  of the filter  380 A and its resonate frequency, f r3 . As shown in  FIG.  21 C  the effective combination of the AW devices  80 A,  80 C,  80 D having the frequency responses  362 A,  362 C,  362 D as shown in  FIG.  21 C  (based on the AW devices physical characteristics) may form the notch filter  380 C with bandwidth  386 C. 
       FIG.  20 A  is a block diagram of a tunable filter module  370 A including electrical elements representing the characteristics of tunable resonators  80 A,  80 B,  80 C according to various embodiments. As shown in  FIG.  20 A , the filter module  370 A may include AW devices  80 A,  80 B,  80 C, variable capacitors  98 A,  98 B, and  98 C, a signal source or generator  92 A, resistors  94 A representing an input load, and a resistor  94 B representing an antenna load. The variable capacitor  98 A may be coupled in parallel to the AW device  80 A. The variable capacitor  98 B may be coupled in parallel to the AW device  80 B. The variable capacitor  98 C may be coupled in series with the AW device  80 C. 
     As shown in  20 A a resonator  80 A,  80 B,  80 C may be represented by a first capacitor  81 A,  81 B,  81 C in parallel with a series coupling of an inductor  86 A,  86 B,  86 C, second capacitor  82 A,  82 B,  82 C, and resistor  84 A,  84 B,  84 C where the capacitors  81 A,  81 B,  81 C,  82 A,  82 B,  82 C may have a capacitance of C OA , C OB , Coc, C MA , C MB , C MC , respectively, inductors  86 A,  86 B,  86 C may have an inductance of L MA , L MB , L MC  and the resistors  84 A,  84 B,  84 C may have a resistance of R MA , R MB , R MC  in an embodiment. As noted the AW devices  80 A,  80 B,  80 C physical characteristics may be selected to create one or filter modules (band-pass  364 C of  FIG.  19 D  and notch  364 D of  FIG.  19 E ). In order for the variable capacitors  98 A,  98 B,  98 C to have a desired tuning effect on the corresponding AW device  80 A,  80 B,  80 C, their capacitance range may need to be significant relative the effective inductance L MA , L MB , L MC  of the AW devices  80 A,  80 B,  80 C. 
     A variable capacitor  98 A,  98 B,  98 C may consume significant die area of a semiconductor including the capacitors and affect the Q (quality) of a filter  370 A including the capacitors  98 A,  98 B,  98 C. In an embodiment a filter  364 D of  FIG.  19 E  may have a center frequency of about 800 MHz. The AW  80 A,  80 B,  80 C may be selected to have resonate frequencies f r1 , f r2 , f r3  of about 797 MHz, 818 MHz, and 800 MHz, respectively. For such a filter the modeled AW devices  80 A,  80 B,  80 C inductance L MA , L MB , L MC  may be about 30 nH, 30 nH, and 132 nH, respectively. In order to effectively tune the AW devices  80 A,  80 B,  80 C, the  98 A,  98 B,  98 C capacitance range may need to be about 4-9.5 pF, 3.5-13 pF, and 2-10 pF in an embodiment. In this example the Q of the resonators may be about 500 and the Q of the variable capacitors  98 A,  98 B, and  98 C may be about 100. 
     In an embodiment, the AW device  80 A may be similar to the AW device  80 B. In this embodiment the variable capacitor  98 A may also be similar to the variable capacitor  98 B. As shown in  FIG.  20 B  a single variable capacitor  98 D may be used to effectively tune both the AW device  80 A and the AW device  80 B. In the filter module  370 B, the variable capacitor  98 D is coupled in parallel to the serial coupled AW devices  80 A, 80. Using the filter module  370 B of  FIG.  20 B , the AW  80 A,  80 B,  80 C may be selected to have resonate frequencies f r1 , f r2 , f r3  of about 800 MHz, 805 MHz, and 800 MHz, respectively. For such a filter the modeled AW devices  80 A,  80 B,  80 C inductance L MA , L MB , L MC  may be about 46 nH, 77 nH, and 44 nH, respectively. In order to effectively tune the AW devices  80 A,  80 B,  80 C of filter  370 B, the  98 D and  98 C capacitance range may need to be about 2-4 pF and 2.5-3.3 pF in an embodiment, a substantial reduction in capacitance relative to the capacitors  98 A,  98 B,  98 C of filter module  370 A of  FIG.  20 A . The filter module or configuration  370 B of  FIG.  20 B  may lower the insertion loss of the filter and improved the Q of the filter module  370 B. In an embodiment, the AW devices  80 A,  80 B, and  80 C may include 41 degree lithium niobate (LiNbO 3 ). 
     As noted above an acoustic wave (AW) device such as  80 A,  80 B,  80 C shown in  FIG.  4   , resonate and anti-resonate frequencies f r0 , f a0  may vary due to manufacturing variants and operating temperature. In addition a variable capacitor such as device such as  98 A,  98 B,  98 C shown in  FIG.  4   , selected or variable capacitance c x0  m (where x is variable capacitance selection x) may vary due to manufacturing variants and operating temperature. In an embodiment, a system such as  430  shown in  FIG.  23    may adjust one or more variable capacitors tuning signals  442 A,  442 B,  442 C based on measured manufacturing variants for AW devices  80 A,  80 B,  80 C and variable capacitors  98 A,  98 B,  98 C and the operating temperature of the system  430  near the AW modules  98 A,  98 B,  98 C. 
     In an embodiment a temperature sensor module  444 A electrically coupled to a contact  444 B near the AW modules  98 A,  98 B,  98 C may calculate the temperature near the AW modules  98 A,  98 B,  98 C. A control logic module  446  may use the calculated temperature and known manufacturing variants for the system  430  components to control or modulate one or more variable capacitors  98 A,  98 B,  98 C via their control signals  442 A,  442 B,  442 C. 
     In an embodiment, the AW modules  98 A,  98 B,  98 C may be configured to operate at a nominal operating temperature where the actual environmental temperature may be below or above the nominal operating temperature. The control logic module  446  may determine the differential between the AW modules’  98 A,  98 B,  98 C nominal operating temperature and the calculated or determined environmental temperature. An AW modules’  98 A,  98 B,  98 C nominal operating temperature may be stored in the PROM  448  ( FIG.  23   ). Further a SPI signal may provide desired settings for the variable capacitors  98 A,  98 B,  98 C. The control logic module  446  may adjust the SPI based settings for the variable capacitors  98 A,  98 B,  98 C based on the calculated environmental temperature and known manufacturing variants for the system  430  components. 
     In an embodiment a programmable read only memory (PROM)  448  may include manufacturing variance characteristics for one or more components  80 A to  80 C and  98 A to  98 C of the system  430 . The PROM  448  characteristics may include the possible resonate and anti-resonate frequencies f r0 , f a0  for each AW module  80 A to  80 C or a delta between the optimal or normal resonate and anti-resonate frequencies f r0 , f a0  and the probable resonate and anti-resonate frequencies f r0 , f a0  for each AW module  80 A to  80 C. The control logic module  446  may use the delta or differential frequency or probable frequency for each AW module  80 A to  80 C to calculate a desired correction to be achieved by modulating a corresponding variable capacitor  98 A to  98 C. 
       FIG.  22 A  is a diagram of a resonant frequency f r0  probably function P r (f)  392 A representing manufacturing variations for an acoustic wave (AW) module according to various embodiments.  FIG.  22 B  is a diagram of an anti-resonant frequency f a0  probably function P a (f)  392 B representing manufacturing variations for an acoustic wave (AW) module according to various embodiments.  FIG.  22 D  is a diagram of a capacitance per unit area c 0  probably function P c (f)  392 D representing manufacturing variations for a capacitor module according to various embodiments. In an embodiment, the PROM  448  may include data representing each P r (f)  392 A, P a (f)  392 B, P c (f)  392 C including the measured standard deviation Δf r0 , Δf a0 , Δf c0  for each function  392 A to  392 C where the functions are approximately Gaussian in nature (as measured or sampled). 
     In an embodiment a programmable read only memory (PROM)  448  may also include temperature variance characteristics for one or more components  80 A to  80 C of the system  430 . The PROM  448  characteristics may include the possible resonate and anti-resonate frequencies f r0 , f a0  for each AW module  80 A to  80 C or a delta between the optimal or normal resonate and anti-resonate frequencies f r0 , f a0  and the probable resonate and anti-resonate frequencies f r0 , f a0  for each AW module  80 A to  80 C based on temperature. The control logic module  446  may use the temperature delta or differential frequency or probable frequency for each AW module  80 A to  80 C to calculate a desired correction to be achieved by modulating a corresponding variable capacitor  98 A to  98 C. 
     In an embodiment, the resonant and anti-resonant frequency variation  392 C for an AW module  80 A to  80 C may be linear as shown in  FIG.  22 C . As shown in  FIG.  22 C  for a positive temperature delta ΔT 0  from a nominal temperature (such as room temperature), an AW module  80 A to  80 C resonant or anti-resonant frequency may be reduced by a predetermined number based on the slope of the temperature function  392 C and magnitude of the temperature delta ΔT 0 . Similarly, as shown in  FIG.  22 C  for a negative temperature delta -ΔT 0  from a nominal temperature (such as room temperature), an AW module  80 A to  80 C resonant or anti-resonant frequency may be increased by a predetermined number based on the slope of the temperature function  392 C and magnitude of the negative temperature delta -ΔT 0 . 
     In an embodiment, the control logic module  446  may combine manufacturing variation deltas and temperature variation deltas provided by the PROM  448  for a component  80 A to  80 C to determine or calculate an overall delta or correction for corresponding variable capacitor  98 A to  98 C. In a further embodiment the control logic module  446  may combine manufacturing variation deltas and temperature variation deltas provided by the PROM  448  for a component  80 A to  80 C and a manufacturing variation deltas provided by the PROM  448  for a corresponding variable capacitor  98 A to  98 C to determine or calculate an overall delta or correction for the corresponding variable capacitor  98 A to  98 C. 
     In an embodiment, the PROM  448  data may be updatable via one or more methods. In such an embodiment the PROM  448  characteristic data for temperature or manufacturing variants for one or more components  80 A to  80 C may be updated based on measured response or updated component testing. Similarly characteristic data for manufacturing variants for one or more capacitors  98 A to  98 C may be updated based on measured response or updated component testing. In an embodiment, the system  430  control logic module  446  may include memory for storing temperature and manufacturing characteristics for components  80 A to  80 C and manufacturing characteristics for components  98 A to  98 C. 
     In order to produce AW modules  80 A to  80 C or variable capacitors  98 A to  98 C or other components having possible variable system characteristics due to manufacturing a process  400  shown in  FIG.  24    may be employed.  FIG.  24    is a flow diagram of a component modeling, manufacturing, and configuration method according to various embodiments. In the process  400  general component characteristics of an AW module  80 A to  80 C or variable capacitor module  98 A to  98 C may be determined. In order to design and manufacture an AW module  80 A to  80 C or variable capacitor module  98 A to  98 C having desired parameters, test devices or related modules may be produced and its characteristics evaluated (activity  402 ). In particular, key or critical parameters may be checked for the test devices including resonant and anti-resonant frequencies for an AW module related device and capacitance per unit area for a capacitor or series of capacitors forming a digital, variable capacitor related device. 
     Based on the test devices and a consistent or well behaved manufacturing process, probability curves or standard deviations for critical parameters of the test devices may be determined. In an embodiment, a Gaussian distribution may be applied and first standard deviations may be determined for each critical parameter probability function. Using correlation(s) between the test devices and an AW module or variable capacitor module to be designed and produced, probability functions (such as each P r (f)  392 A, P a (f)  392 B, P c (f)  392 C) may be determined for the AW module or variable capacitor modules. 
     Based on the correlations between the test devices and resultant probability functions for critical parameters, an AW module or capacitor module may be designed (activity  404 ). Without compensating modules or methods as recited by the present invention, an AW module or capacitor module design parameters may be required to be loose to compensate for the manufacturing variants. Employing the AW modules or capacitors in a system  430  (with compensating modules) of the present invention may enable tighter design parameters given the ability to compensate for variants of the system  430 . In an embodiment initial, final components (AW module or capacitor modules) based on a design may be produced (activity  406 ). Then, the initial components based on the associated design may be tested to determine the probability characteristics for key or critical parameters (activity  408 ). 
     The determined probability characteristics for the initial final, designed components may be compared to the determined probability characteristics for the test devices. Where the characteristics are correlated as expected, larger quantities of the final, design components may be produced and randomly tested (activity  412 ). Where the manufacturing process and source is controlled and well-behaved only sparse or random components may need to be tested to confirm correlation to the previously determined probability functions P r (f)  392 A, P a (f)  392 B, P c (f)  392 C. For temperature sensitive components including AW modules, the temperature effects may also be modeled (activity  402 ) and considered during the component design (activity  404 ). The temperature characteristics of initial, final components may also be determined (activity  408 ) prior to producing higher quantities of temperature sensitive components (activity412). In an embodiment each or batch groups of final, designed component (AW module or variable capacitor module) may be tested and resultant probability function determined for key or critical module characteristics. As noted the determined probability functions may be stored in a system  430  employing a corresponding module ( 80 A to  80 C,  98 A to  98 C). 
     In addition to adjusting for AW modules’ performance variants due manufacturing variants and operating temperature, impedances present at a filter module  452 A input or output port may affect the filter module  452 A ( FIG.  25 A ) performance. In particular a filter module  452 A may be designed for a particular load at its input node and a particular load at its output node. In an embodiment a differential between the target/designed load  94 A on the input node or the target/designed load  94 B on the output node of a filter module  452 A may affect its performance.  FIG.  25 A  is block diagram of signal filter architecture  450 A. Architecture  450 A includes a filter module  452 A, an input load  94 A represented by a resistor and an output load  94 B represented by a resistor. The filter module  452 A may be configured to have a balanced load where the input load impedance  94 A and the output load impedance  94 B are about equal and have a predetermined level such as  50  ohms in an embodiment. 
     The ratio between target loads  94 A,  94 B is related to the Voltage Standing Wave Ratio (VSWR) for the module. As noted, a filter module  452 A may be configured for a common VSWR of 1:1 (where the input load  94 A is about equal to the output load  94 B). For a filter module  452 A configured for a VSWR of 1:1 an input-output mismatch (VSWR other than 1:1) may result in a greater input signal insertion loss (greater filter passband loss).  FIG.  25 B  is a block diagram of a signal filter architecture  450 B including a tunable filter module  452 B that may be configured to reduce effects of impedance mismatches between loads  94 A,  94 B (VSWR other than expected by filter module  452 A,  452 B nominally). 
     As shown in the  FIG.  25 B  the signal filter architecture  450 B includes an input load  94 A, an output load  94 B, and a tunable filter module  452 B. The tunable filter module  452 B includes multiple tunable AW modules  96 A,  96 C,  96 D,  96 E. Each tunable AW module  96 A,  96 C,  96 D,  96 E may include an AW device  80 A,  80 C,  80 D,  80 E, and  80 F (represented by their electrical component equivalents) coupled in parallel to a variable capacitor  98 A,  98 C,  98 D,  98 E, and  98 F, respectively. The tunable AW module  96 C may be coupled to the input load  94 A and ground. One or more sub-filter modules  454 A,  454 B may be coupled between the tunable AW module  96 C and the output load  94 B. 
     Each sub-filter module  454 A,  454 B may include a first tunable AW module  96 C,  96 E and a second tunable AW module  96 D,  96 F coupled to ground, respectively. As noted above an AW device  80 A,  80 C,  80 D,  80 E,  80 F may be modeled from a series of a inductor  86 A,  86 C,  86 D,  86 E,  86 F, capacitor  82 A,  82 C,  82 D,  82 E,  82 F, resistor  84 A,  84 C,  84 D,  84 E,  84 F coupled in parallel with a capacitor  81 A,  81 C,  81 D,  81 E,  81 F, respectively. Each variable capacitor  98 A,  98 C,  98 D,  98 E, and  98 F coupled in parallel with an AW device  80 A,  80 C,  80 D,  80 E, and  80 F may be varied to affect the filter characteristics of the AW device  80 A,  80 C,  80 D,  80 E, and  80 F. 
     As noted previously a variable capacitor  98 A,  98 C,  98 D,  98 E, and  98 F may be employed to modulate an AW device  80 A,  80 C,  80 D,  80 E, and  80 F to shift a resonant or anti-resonant frequency to select different bands, sub-bands, correct for manufacturing variants, and temperature shifts. A variable capacitor  98 A,  98 C,  98 D,  98 E, and  98 F may also be employed to modulate an AW device  80 A,  80 C,  80 D,  80 E, and  80 F to reduce a input signal insertion loss due to an unexpected or non-conforming VSWR (not equal to VSWR the filter model  452 B was designed to process). 
     In an embodiment, the filter module  452 B may be designed for a VSWR of about 1:1 and the variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F may be modulated to reduce insertion loss due to a VSWR other than 1:1 (non-forming). For example,  FIG.  26 A  is a diagram of the frequency response of the filter module  452 B for a VSWR of 1:1 (nominal). As shown in  FIG.  26 A  the insertion loss (passband attenuation) is about 0.5 dB.  FIG.  26 B  is a diagram of the frequency response of the filter module  452 B for a VSWR of 1:1.5 and one or more variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F modulating a AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively to reduce the insertion loss. As shown in  FIG.  26 B  the insertion loss (passband attenuation) is about 0.68 dB.  FIG.  26 C  is a diagram of the frequency response of the filter module  452 B for a VSWR of 1:2 and one or more variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F modulating a AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively to reduce the insertion loss. As shown in  FIG.  26 C  the insertion loss (passband attenuation) is about 1 dB. 
     In another embodiment the PROM  448  of  FIG.  23    may be configured to include variable capacitor deltas for various VSWR. A user may be indicate the output load and configure the PROM  448  accordingly. In another embodiment the control logic module may sense the output load, determine the VSWR differential, and choose the closest set of variable capacitor deltas from the PROM  448 . In a further embodiment a filter module  452 B may be configured or designed for a nominal VSWR (median relative to possible VSWR that the filter module  452 B may experience). For example in architecture  450 B, VSWRs of 1:1, 1:1.5 and 1:2 may be expected. The filter module  452 B may be configured or designed to be optimal for a VSWR of 1:1.5 and the variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F may be adjusted to modulate the AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively when the VSWR is 1:1 or 1:2. In a further embodiment a variable capacitor may be placed in series with a AW module  80 C,  80 F (or  80 A,  80 E) (such as capacitor  98 C in  FIG.  20 A ). The variable capacitor in series with an AW module  80 C,  80 F may be modulated to compensate for loads  94 A,  94 B other than the target/designed loads of the filter module  450 B. 
       FIG.  27 A  is a diagram of the frequency response of the filter module  452 A for a VSWR of 1:1 where the filter module  452 B is optimized for VSWR of 1:1, 1:1.5, and 1:2 and one or more variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F modulate a AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively to reduce the insertion loss for VSWR 1:1. As shown in  FIG.  27 A  the insertion loss (passband attenuation) is about 0.65 dB.  FIG.  27 B  is a diagram of the frequency response of the filter module  452 B for a VSWR of 1:1.5 where the filter module  452 A is optimized for VSWR of 1:1, 1:1.5, and 1:2 and one or more variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F modulate a AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively to reduce the insertion loss for VSWR 1:1.5. As shown in  FIG.  27 B  the insertion loss (passband attenuation) is about 0.62 dB.  FIG.  27 C  is a diagram of the frequency response of the filter module  452 B for a VSWR of 1:2 where the filter module  452 B is optimized for VSWR of 1:1, 1:1.5, and 1:2 and one or more variable capacitors  98 A,  98 C,  98 D,  98 E, and  98 F modulate a AW device  80 A,  80 C,  80 D,  80 E, and  80 F, respectively to reduce the insertion loss for VSWR 1:2. As shown in  FIG.  27 C  the insertion loss (passband attenuation) is about 0.69 dB. 
     As shown in  FIGS.  26 A to  26 C  the average insertion loss is about 0.72 dB for a system designed for a VSWR 1:1 and adjusted for VSWR of 1:1.5 and 1:2. As shown in  FIGS.  27 A to  27 C  the average insertion loss is about 0.65 dB for a system optimized for a range of VSWR from 1:1 to 1:2 and adjusted for VSWR of 1:1.0, 1:1.5, and 1:2. The insertion loss of the filter module  452 B optimized for VSWR 1:1 has a lower insertion loss for VSWR 1:1 than the insertion loss for the filter module  452 B optimized for a range of VSWR from 1:1 to 1:2 (0.5 dB versus 0.65 db) even with variable capacitor modulation. Accordingly different filter modules  452 B for VSWR 1:1 optimization or a range of VSWR may be selected as a function of the expected range of VSWR in a system implementation and minimal acceptable insertion loss criteria. 
     As noted the VSWR is based on the balance between the input load and output load of a system. As shown in  FIG.  1 A  and  FIG.  28 A , a power amplifier  12  may, in part provide a load to filter module  452 A ( FIG.  28 A ). Power amplifiers  12  commonly produce very low impedance. In order to provide a desired input impedance to the filter module  452 A ( FIG.  28   ) or RF switch  40  ( FIG.  1 A ), one or more elements forming an impedance matching module  470 A may be placed between the PA  12  and filter module  462 A. The impedance matching module  470 A may provide the expected impedance at the input port of a filter module  462 A. When the filter module  462 A is tunable and support filtering different frequency bands, the matching module  470 A may not be effective for all the various operating/filtering modes of the tunable filter module  462 A. 
       FIG.  28 A  is a block diagram of a filter system architecture  460 A according to various embodiments. Architecture  460 A includes a PA  12 , an impedance matching module  470 A and a tunable/switchable filter module  462 A. The impedance matching module  470 A couples the PA  12  to the tunable/switchable filter module  462 A. In an embodiment, the tunable/switchable filter module  462 A includes a variable capacitor control signal SPI and a band select signal. The tunable/switchable filter module  462 A may produce or switch between different frequency responses to process different frequency spectrum or bands. In an embodiment, the impedance matching module  470 A may include an inductor  464 A. The PA  12  may receive power via input VDD in an embodiment. 
     The inductor  464 A may provide the impedance matching function of the impedance matching module  470 A. In an embodiment, the inductor may be about a 2 to 3 nH inductor.  FIG.  28 B  is a block diagram of a tunable/switchable signal filter module  462 B that may be configured to operate in multiple bands and provide impedance matching with the matching module  470 A. In an embodiment, the filter module  462 B may be configured to operate in evolved UMTS Terrestrial Radio Access Network e-UTRAN Long Term Evolution (LTE) bands, in particular bands  13  and  17 . LTE band  13  may have a transmit band from 776 MHz to 787 MHz and a receive band from 746 MHz to 757 MHz. LTE band  17  may have a transmit band from 704 MHz to 716 MHz and a receive band from 734 MHz to 746 MHz. LTE Bands  13  and  17  are adjacent, tight bands. 
     As shown in the  FIG.  28 B  tunable/switchable filter module  462 B includes multiple tunable AW modules  476 C,  476 D,  476 F and multiple tunable/switchable AW modules  476 A,  476 E. Tunable AW module  476 C may include AW devices  80 C and  80 E coupled in parallel, the set coupled in parallel to a variable capacitor  98 C. The tunable AW module  476 C may be coupled to the impedance matching module  470 A and ground. Tunable AW module  476 D,  476 F may include an AW device  80 D,  80 G coupled in parallel to a variable capacitor  98 D,  98 F, respectively. One or more sub-filter modules  474 A,  474 B may be coupled between the tunable AW module  96 C and the output load  94 B. 
     Each sub-filter module  474 A,  474 B may include a first tunable/switchable AW module  476 A,  476 E and a second tunable AW module  476 D,  476 F coupled to ground, respectively. Tunable AW module  476 A may include AW device  80 A in series with a switch  472 B coupled in parallel to AW device  80 F in series with a switch  472 A, the set coupled in parallel to a variable capacitor  98 A. Tunable AW module  476 E may include AW device  80 H in series with a switch  472 C coupled in parallel to AW device  80 I in series with a switch  472 D, the set coupled in parallel to a variable capacitor  98 E. 
     In a first mode the switches  474 A to  474 D may operate to switch AW module  80 A and AW module  80 H on (closed) and AW module  80 F and AW module  80 I off (switch open) for band  13  or  17 . In a second mode the switches  474 A to  474 D may operate to switch AW module  80 A and AW module  80 H off (switch open) and AW module  80 F and AW module  80 I on or active (switch closed) for the other of band  13  or  17 . The variable capacitors  98 A,  98 E,  98 F, and  98 D may be employed to adjust the operation of the AW modules  80 F,  80 A,  80 I,  80 H,  80 G, and  80 D to correct for temperature, output impedance, and manufacturing variants. It is noted that variable capacitor  98 A modulates AW module  80 A or  80 F (is shared) and variable capacitor  98 E modulates AW module  80 H or  80 I (is shared). 
     The variable capacitor  98 C may be modulated to provide impedance matching between the filter module  462 B and the impedance matching module  470 A.  FIG.  29 A  is a diagram of the frequency response of the tunable/switchable filter module  462 B operating in a first mode to pass signals for LTE band  17  in an embodiment.  FIG.  29 B  is a diagram of the frequency response of the tunable/switchable filter module  462 B operating in a second mode to pass signals for LTE band  13  in an embodiment. In an embodiment, the parallel combination of AW modules  80 C and  80 E are configured to resonate about the LTE band  17  and thereby provide rejection below LTE band  17  and between LTE band  17  and  13 . The variable capacitor  98 C may also tune the anti-resonant point between LTE band  17  and  13  as a function of the mode of operation (mode 1 or mode 2). 
     In an embodiment, the switches  472 A to  472 D may be comprised of stacked CMOS FETs to pass the PA amplified signals. The use of multiple sub-filters  474 A,  474 B in series may reduce the stack size and power across the switches  474 A to  474 D as the signal is shared across the sub-filters. In a further embodiment the capacitors  98 A and  98 E may be fixed. Their capacitance may be preset based on known manufacturing variants, operating temperature variants, and impedance matching (output) corrections that are fixed for the filter module  462 B. In another embodiment of all the variable capacitors  98 A to  98 G described in the application capacitance range and granularity may be varied as function of corrections needed to maintain the associated AW modules  80 A to  80 G nominal resonant and anti-resonant frequencies within acceptable tolerances. The corrections may be known or calculated based on the AW modules  80 A to  80 G known manufacturing and operating temperature variants and output impedance compensation conditions. 
       FIG.  30 A  is a simplified block diagram of a signal filter architecture according to various embodiments. As shown in  FIG.  30 A  signal filter architecture  480 A includes a source  92 A, a resistor  94 A, a signal processing module  482 A, and a resistor  94 B. The resistor  94 A may represent the input load generated by the signal  92 A. The signal processing module  482 A may modify or filter the source signal  92 A in a desired or predetermined way. The second resistor  94 B may represent the load at an output including at an antenna. As noted, a differential between the target/designed load or impedance  94 A on the input node or the target/designed load  94 B on the output node of a filter module  480 A may affect its performance. It is noted that the loads  94 A,  94 B may have real and imaginary components in an embodiment (x + jy) where x is the real component and y is the imaginary component. 
     As noted, the ratio between loads or impedance  94 A,  94 B is related to the Voltage Standing Wave Ratio (VSWR) for the module  480 A where a module  480 A may be configured for a common VSWR of 1:1 (where the input impedance  94 A is about equal to the output impedance  94 B). For a filter module  482 A configured for a VSWR of 1:1 an input-output mismatch (VSWR other than 1:1) may increase an input signal  92 A insertion loss (greater filter passband loss). 
       FIG.  30 B  is a simplified block diagram of an impedance matched (“IM”) signal filter architecture  480 B according to various embodiments. Architecture  480 B includes a signal source  92 A, a resistor  94 A, a signal processing module  482 A, a resistor  94 B, and an impedance match (“IM”) module  484 A. In an embodiment, the IM module  484 A may include one or more components that are selected or configured to provide a real or imaginary impedance balance, modification, or modulation between input and output impedance for architecture  480 A. In an embodiment, the IM module response or modulation may vary by frequency and thus be tuned for a range or ranges of desired frequencies such as when frequency variable components are employed in an IM module  484 A such as shown  FIG.  30 C . 
       FIG.  30 C  is a simplified block diagram of an impedance matched (“IM”) signal filter architecture  480 C including an IM module  484 C according to various embodiments. As shown in  FIG.  30 C , architecture  480 C includes a signal source  92 A, a resistor  94 A, a signal processing module  482 A, a resistor  94 B, and an impedance match (“IM”) module  484 C. In an embodiment, a IM module  484 C may include one or more frequency variant components that are selected or configured to provide a real or imaginary impedance balance, modification, or modulation between input and output impedance for architecture  480 C. In an embodiment, the IM module  484 C may include an L-shaped resonator circuit. The module  484 C may include an inductor  86 J coupled serially between the signal processing module (SPM)  482 A and the resistor  94 B. The module  484 C may further include a capacitor  82 J coupled between the inductor  86 J and resistor  94 B and ground. 
     The resultant L-C circuit formed by the inductor  86 J and the capacitor  82 J may provide balancing impedance between the source  94 A and output port  94 B. As a function of the inductance and capacitance of the inductor  86 J and capacitor  82 J and loads  94 A,  94 B, the L-C circuit of module  484 C may balance the impedance  94 A,  94 B at or about predetermined frequenc(ies). When the input impedance  94 A is about 50 ohms and the output impedance  94 B is about 100 ohms, the VSWR may be about 1:2 causing about a 6 dB insertion loss for an input signal  92 A. In an embodiment, the inductance and capacitance of the inductor  86 J and capacitor  82 J may be about 9.406 µH (micro-Henries) and 1.881 pF (pico-Farads). In such an embodiment, the IMM  484 C may provide an impedance of about 50 ohms about a frequency of 846 MHz. In this embodiment, the IMM  484 C may balance the source and output impedance so the VSWR is about 1:1 and the input signal insertion loss about 2 dB. 
     It is noted that the inductor  86 J may consume substantial real estate and lower the quality (Q) of architecture  480 C due to its substantial inductance. In an embodiment, it may be desirable to balance architecture impedance while not employing a large inductor as in an L-C resonator circuit shown in  FIG.  30 C  or variants thereof including T-networks ( 484 N in  FIG.  30 T ).  FIG.  30 D  is a simplified block diagram of IM signal filter architecture  480 D including an IMM  484 D. Similar to architecture  480 C, architecture  480 D includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, a SPM  482 , an IMM  484 D, and an output load or impedance represented by resistor  94 B (such as an antenna impedance in an embodiment). In an embodiment, the IMM  484 D may include an acoustic wave module (AWM)  490 A, a pre-impedance match component module (pre-IMCM)  491 A, and a post-impedance match component module (post-IMCM)  492 A. The AWM  490 A may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. 
     In an embodiment, a post-IMCM  492 A may include one or more components configured along with the AWM  490 A to create a balancing impedance between  94 A and  94 B, such as  492 B in  FIGS.  30 F,  492 C  in  FIGS.  30 I, and  492 D  in  FIG.  30 R . Similarly, a pre-IMCM  491 A may include one or more components configured along with the AWM  490 A to create a balancing impedance (real and imaginary) between  94 A and  94 B, such as  492 B in  FIGS.  30 L and  492 C  in  FIG.  30 O . In an embodiment, one or more components of a post-IMCM  492 A may be configured to resonate with an AWM  490 A to add impedance to architecture  480 D, i.e., when impedance of  94 B &gt;  94 A. In an embodiment, one or more components of a pre-IMCM  491 A may be configured to resonate with an AWM  490 A to reduce the impedance of architecture  480 D, i.e., when impedance of  94 A &gt;  94 B. 
     In other embodiments, an IMM’s  484 D pre-IMCM  491 A and post-IMCM  492 A may both include one or more components configured to interact or resonate with the AWM  490 A to affect architecture  480 D input/output impedance ratios or VSWR. In an embodiment, the AWM  490 A of a IMM  484 D may be configured to filter an input signal  92 A in addition to resonating with one or more components of a pre-IMCM  491 A or post-IMCM  492 A to modulate or modify the impedance ratio of architecture  480 D for various frequencies. As a function of the pre-IMCM  491 A and post-IMCM  492 A components the resonate frequency, f r  and the anti-resonance f a  of the AWM  490 A may be shifted or modified in a predetermined and configurable manner. In particular, the AWM nominal resonate frequency, f r  and the anti-resonance f a  may be selected based on the known shift of these frequencies due to the interaction with components of a pre-IMCM  491 A or post-IMCM  492 A. 
       FIG.  30 E  is a simplified block diagram of IM architecture  480 E including an IMM  484 D according to various embodiments. Similar to architecture  480 D, architecture  480 E includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 D, and an output load or impedance represented by resistor  94 B (such as an antenna impedance in an embodiment). In an embodiment, the IMM may include an acoustic wave module (AWM)  490 A, a pre-impedance match component module (pre-IMCM)  491 A, and a post-impedance match component module (post-IMCM)  492 A. The AWM  490 A may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. Architecture  480 E may not include a SPM  482  as shown in  FIG.  30 D . It is noted that an IMM  484 D of the present invention may be employed in various networks or architecture to modify the architecture impedance ratio in a known or desired way. 
       FIG.  30 F  is a simplified block diagram of IM architecture  480 F including an IMM  484 F according to various embodiments. Similar to architecture  480 E, architecture  480 F includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 F, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 F may include an acoustic wave module (AWM)  490 A and a post-impedance match component module (post-IMCM)  492 B. The AWM  490 A may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. In  FIG.  30 F , the AWM  490 A is shown as a single AW device with representative electrical components or elements including capacitors  81 A,  82 A, inductor  86 A, and resistor  84 A, the capacitors  81 A,  82 A having capacitance Cr and Cm, respectively, the inductor  86 A inductance Lm, and resistor 84 resistance Rm. 
     In an embodiment, the IMM  484 F may be configured to provide a balancing impedance to architecture  480 F at a desired or target frequency. The IMM  484 F may be configured to add impedance to architecture  480 F when output impedance is greater than the input impedance. In IMM  484 F, a capacitor  82 J is coupled to ground and between the AWM  490 A and the output load  94 B to form a resonator circuit with the AWM  490 A, the resonator circuit having a desired impedance at desired frequency f r&#39; . The AWM  490 A may be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the capacitor  82 J where the capacitor  82 J effectively borrows inductance L (represented by inductor  86 J in  FIGS.  30 G and  30 H ) from the AWM  490 A to resonate at the desired frequency f r&#39;  and provide the desired impedance at that the desired frequency f r&#39; . 
     As shown in  FIG.  30 H , the AWM  490 A provides an inductor  86 J having inductance L to the L-C module  484 C. The AWM  490 A is then effectively coupled to an inductor  86 K with inductance -L forming the modified AWM  494 A (balanced inductance as shown by module  486 A in  FIG.  30 G ). In  FIG.  30 G , the balanced inductor pair module  486 A includes an inductor  86 K having inductance -L and an inductor  86 J having inductance L. The module  486 A represents the effect of the AWM  490 A losing inductance L (inductor  86 K) to the capacitor  82 J so the capacitor  82 J and effective inductor  86 J having inductance L form a desired resonator module  484 C. The balanced inductor pair module  486 A represents the net effect of capacitor  82 J resonating with AWM 490A: the AWM  490 A providing inductance L to resonate with the capacitor  82 J (L-C module  484 C of  FIG.  30 H ), while the AWM  490 A losses inductance L. The AWM  490 A resonance may be adjusted accordingly (based on modified AWM  494 A of  FIG.  30 H ). 
     Using the values Cr, Cm, Lm, and Rm for first capacitor  82 A, inductor  86 A, second capacitor  82 B, and resistor  84 A, a AWM’s  490 A nominal resonance frequency f r  may be defined by the following equation: f r  ≡  
     
       
         
           
             
               1 
               
                 2 
                 π 
                 
                   
                     
                       L 
                       m 
                     
                     
                       C 
                       m 
                     
                   
                 
               
             
             . 
           
         
       
     
     In an embodiment, the capacitance of Cr is modified so the modified resonator  494 A (with -L) resonates at f r&#39; . Cr may be determined when  
     
       
         
           
             w 
             
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               r 
             
             
                 
               2 
             
             &gt; 
             
               
                 L 
                 m 
               
               L 
             
             
               
                 w 
                 
                   &#39; 
                   r 
                 
                 
                     
                   2 
                 
                 − 
                 
                   w 
                   r 
                 
                 
                     
                   2 
                 
               
             
           
         
       
     
     and f r&#39;  &gt; f r , then Cr =  
     
       
         
           
             
               
                 L 
                 ⋅ 
                 w 
                 
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                   r 
                 
                 
                     
                   2 
                 
                 + 
                 L 
                 m 
                 ⋅ 
                 
                   
                     w 
                     
                       &#39; 
                       r 
                     
                     
                         
                       2 
                     
                     − 
                     
                       w 
                       r 
                     
                     
                         
                       2 
                     
                   
                 
               
               
                 L 
                 ⋅ 
                 L 
                 m 
                 ⋅ 
                 w 
                 
                   &#39; 
                   r 
                 
                 
                     
                   2 
                 
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                     w 
                     
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                       r 
                     
                     
                         
                       2 
                     
                     − 
                     
                       w 
                       r 
                     
                     
                         
                       2 
                     
                   
                 
               
             
             ⋅ 
           
         
       
     
     Accordingly based on desired effective L-R module  484 C having impedance at f r&#39; , a AWM  490 A may be configured to provide inductance L and resonate at f r&#39; . 
     In an embodiment, IMM  484 F may be configured to balance an input impedance  94 A of about 50 ohms with an output impedance  94 B about 100 ohms for a frequency f r&#39;  of 846 MHz. Similar to the capacitor  82 J of L-C module  484 C of  FIG.  30 C , capacitor  82 J may have a capacitance of about 1.881 pF (pico-Farads). In the L-C module  484 C of  FIG.  30 C , the inductor  86 J had an inductance of about 9.406 nH (nano-Henries) where the L-C module  484 C provided the desired impedance at 846 MHz. Accordingly the AWM  490 A may be provide an effective inductance of 9.406 nH (nano-Henries) to the L-C module  484 C shown in  FIG.  30 H  and lose the same inductance to form the modified AWM  494 A shown in  FIG.  30 H . In an embodiment, Lm may be about 100 nH (nano-Henries) and Cm may be about 0.3713 pF (pico-Farads). Using the equation for Cr above, Cr may be about 3.813 pF (pico-Farads) where the AWM  490 A has a nominal frequency of 826 MHz. 
     The IMM  484 F may provide an impedance of about 50 ohms about a frequency of 846 MHz. In this embodiment, the IMM  484 F may balance the source and output impedance so the VSWR is about 1:1 and the input signal insertion loss is nominal as shown in the frequency response graph  498 B in  FIG.  31 B .  FIG.  31 B  is a frequency response graph  498 A of the AWM  490 A alone with a balanced load applied to the AWM  490 A. As shown in  FIG.  31 A , the AWM  490 A has a nominal resonate frequency f r  of about 826 MHz. When the AWM  490 A is coupled with the capacitor  82 J in architecture  480 F having an unbalanced input-output impedance, (1:2), the IMM  484 F (AWM  490 A and capacitor  82 J) may combine to have a resonate frequency f r&#39;  of about 846 MHz as shown in  FIG.  31 B . As shown in Table 1 different AWM and capacitances for capacitor  82 J may be employed to achieve a resonate frequency of about 836 MHz and various impedances (resistance from about 82 ohms to 413 ohms.) 
     
       
         
          TABLE 1
           
               
               
               
               
               
               
               
               
               
               
             
               
                 Fr 
                 Fa 
                 F′r 
                 Lm 
                 Cm 
                 Cr 
                 L 
                 C 
                 R 
                 Q 
               
             
            
               
                 MHz 
                 MHz 
                 MHz 
                 n-H, 
                 p-F, 
                 p-F 
                 n-H 
                 p-F 
                 Ohms 
                   
               
               
                 750 
                 800 
                 836 
                 30 
                 1.501 
                 10.89 
                 7.704 
                 1.862 
                 82.75 
                 0.8 
               
               
                 750 
                 800 
                 836 
                 60 
                 0.7501 
                 5.447 
                 15.41, 
                 1.702 
                 181.01 
                 1.6 
               
               
                 750 
                 800 
                 836 
                 100 
                 0.4503 
                 3.268 
                 25.68 
                 1.241 
                 413.93 
                 2.4 
               
            
           
         
       
     
     Other components may be coupled with an AWM in an embodiment for various desired impedance matches at various desired resonate frequencies.  FIG.  30 I  is a simplified block diagram of IM architecture  480 I including an IMM  484 G according to various embodiments. Similar to architecture  480 F, architecture  480 I includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 G, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 G may include an acoustic wave module (AWM)  490 B and a post-impedance match component module (post-IMCM)  492 C. The AWM  490 B may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. As shown in  FIG.  30 F , the AWM  490 B may be described with representative electrical components or elements including capacitors  81 A,  82 A, inductor  86 A, and resistor  84 A, the capacitors  81 A,  82 A having capacitance Cr and Cm, respectively, the inductor  86 A inductance Lm, and resistor 84 resistance Rm. 
     In an embodiment, the IMM  484 G may be configured to provide a balancing impedance to architecture  480 I at a desired or target frequency. The IMM  484 G may be configured to add impedance to architecture  480 G when output impedance is greater than the input impedance. In IMM  484 G, an inductor  86 L is coupled to ground and between the AWM  490 B and the output load  94 B to form a resonator circuit with the AWM  490 B, the resonator circuit having a desired impedance at desired frequency f r&#39; . The AWM  490 B may be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the inductor  86 L where the inductor  86 L effectively borrows capacitance C (represented by capacitor  82 K in  FIGS.  30 J and  30 K ) from the AWM  490 B to resonate at the desired frequency f r&#39;  and provide the desired impedance at that the desired frequency f r&#39; . 
     As shown in  FIG.  30 I , the AWM  490 B provides a capacitor  82 K having capacitance C to the L-C module  484 H. The AWM  490 B is then effectively coupled to a capacitor  82 L with capacitance -C forming the modified AWM  494 B (balanced capacitance is shown by module  486 B in  FIG.  30 J ). In  FIG.  30 J , the balanced capacitor pair module  486 B includes a capacitor  82 K having capacitance -C and a capacitor  82 L having capacitance -C. The module  486 B represents the effect of the AWM  490 B losing capacitance C (capacitor  82 K) to the inductor  86 L so the inductor  86 L and effective capacitor  82 K having capacitance C form a desired resonator module  484 G. The balanced capacitor pair module  486 B represents the net effect of inductor  86 L resonating with AWM 490B: the AWM  490 B providing capacitance C to resonate with the inductor  86 L (L-C module  484 H of  FIG.  30 K ), while the AWM  490 B losses capacitance C. The AWM  490 B resonance may be adjusted accordingly (based on modified AWM  494 B of  FIG.  30 K ). 
     In an embodiment, IMM  484 G may be configured to balance an input impedance  94 A of about 50 ohms with an output impedance  94 B about 414 ohms for a frequency f r&#39;  of 846 MHz. The inductor  89 L may have an inductance of about 29.21 nH. The capacitor 82 K may have an effective capacitance of about 1.411 pF and the capacitor  82 L may have an effective capacitance of about -1.411 pF. Accordingly, the AWM  490 A may provide an effective capacitance of about 1.411 pF to the L-C module  484 H shown in  FIG.  30 K  and lose the same capacitance to form the modified AWM  494 B shown in  FIG.  30 K . In an embodiment, Lm may be about 100 nH (nano-Henries), Cm may be about 0.4503 pF (pico-Farads), and Cr may be about 3.268 pF (pico-Farads). 
     The IMM  484 G may provide an impedance of about 364 ohms about a frequency of 836 MHz. In this embodiment, the IMM  484 G may balance the source and output impedance so the VSWR is about 1:1 and the input signal insertion loss is nominal as shown in the frequency response graph  498 C in  FIG.  31 C . When the AWM  490 B is coupled with the inductor  86 L in architecture  480 I, the IMM  484 G (AWM  490 B and inductor  86 L) combine to have a resonate frequency f r&#39;  of about 836 MHz as shown in  FIG.  31 C . 
       FIGS.  30 L to  30 N  are simplified diagrams of another embodiment  480 L that includes an IMM  484 I. Similar to architecture  480 F, architecture  480 L includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 I, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 I may include an acoustic wave module (AWM)  490 A and a pre-impedance match component module (pre-IMCM)  491 B. The AWM  490 A may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. In  FIGS.  30 L- 30 N , the AWM  490 A is shown as a single AW device but may be represented by electrical components or elements including capacitors  81 A,  82 A, inductor  86 A, and resistor  84 A, the capacitors  81 A,  82 A having capacitance Cr and Cm, respectively, the inductor  86 A inductance Lm, and resistor 84 resistance Rm. 
     In an embodiment, the IMM  484 I may be configured to provide a balancing impedance to architecture  480 L at a desired or target frequency. The IMM  484 I may be configured to remove impedance from architecture  480 L when output impedance is less than the input impedance. In IMM  484 I, a capacitor  82 J is coupled to ground and between the AWM  490 A and the input load  94 A to form a resonator circuit with the AWM  490 A, the resonator circuit having a desired impedance at desired frequency f r&#39; . The AWM  490 A may be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the capacitor  82 J where the capacitor  82 J effectively borrows inductance L (represented by inductor  86 J in  FIGS.  30 M and  30 N ) from the AWM  490 A to resonate at the desired frequency f r&#39;  and provide the desired impedance at that the desired frequency f r&#39; . As shown in  FIG.  30 N , the AWM  490 A provides an inductor  86 J having inductance L to the L-C module  484 J. The AWM  490 A is then effectively coupled to an inductor  86 K with inductance -L forming the modified AWM  494 C (balanced inductance as shown by module  486 C in  FIG.  30 M ). 
       FIGS.  30 O to  30 Q  are simplified diagrams of another embodiment  480 M that includes an IMM  484 K. Similar to architecture  480 L, architecture  480 M includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 K, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 K may include an acoustic wave module (AWM)  490 B and a pre-impedance match component module (pre-IMCM)  491 C. The AWM  490 B may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. In  FIGS.  30 O- 30 Q , the AWM  490 B is shown as a single AW device but may be represented by electrical components or elements including capacitors  81 A,  82 A, inductor  86 A, and resistor  84 A, the capacitors  81 A,  82 A having capacitance Cr and Cm, respectively, the inductor  86 A inductance Lm, and resistor 84 resistance Rm. 
     In an embodiment, the IMM  484 K may be configured to provide a balancing impedance to architecture  480 M at a desired or target frequency. The IMM  484 K may be configured to remove impedance from architecture  480 M when output impedance is less than the input impedance. In IMM  484 K, an inductor  86 L is coupled to ground and between the AWM  490 B and the input load  94 A to form a resonator circuit with the AWM  490 B, the resonator circuit having a desired impedance at desired frequency f r&#39; . The AWM  490 B may be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the inductor  86 L where the inductor  86 L effectively borrows capacitance C (represented by capacitor  82 K in  FIGS.  30 M and  30 N ) from the AWM  490 B to resonate at the desired frequency f r&#39;  and provide the desired impedance at that the desired frequency f r&#39; . As shown in  FIG.  30 Q , the AWM  490 B may provide a capacitor  82 K having capacitance C to the L-C module  484 L. The AWM  490 B is then effectively coupled to a capacitor  82 L with capacitance -C forming the modified AWM  494 D (balanced capacitance as shown by module  486 D in  FIG.  30 P ). 
       FIGS.  30 R to  30 T  are simplified diagrams of another embodiment  480 N that includes an IMM  484 M. Similar to architecture  480 F, architecture  480 N includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 M, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 M may include an acoustic wave module (AWM)  490 A and a post-impedance match component module (post-IMCM)  492 D. The AWM  490 A may be a single acoustic wave device or a plurality of devices and variable capacitors in various configurations as shown and described above. In  FIGS.  30 R- 30 T , the AWM  490 A is shown as a single AW device but may be represented by electrical components or elements including capacitors  81 A,  82 A, inductor  86 A, and resistor  84 A, the capacitors  81 A,  82 A having capacitance Cr and Cm, respectively, the inductor  86 A inductance Lm, and resistor 84 resistance Rm. 
     In an embodiment, the IMM  484 M may be configured to provide a balancing impedance to architecture  480 N at a desired or target frequency. The IMM  484 M may be configured to add impedance to architecture  480 N when output impedance is greater than the input impedance. In IMM  484 M, the post-IMCM  492 D includes a capacitor  82 J and inductor  86 N. The capacitor  82 J is coupled to ground and between the AWM  490 A and the inductor  86 N. The inductor  86 N is coupled between the AWM  490 A and the output load  94 B. The post-IMCM  492 D forms a T-shaped resonator circuit with the AWM  490 A, the resonator circuit having a desired impedance at desired frequency f r&#39; . The AWM  490 A may be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the capacitor  82 J where the capacitor  82 J effectively borrows inductance L (represented by inductor  86 J in  FIGS.  30 S and  30 T ) from the AWM  490 A to resonate at the desired frequency f r&#39;  and provides the desired impedance at that the desired frequency f r&#39; . As shown in  FIG.  30 T , the AWM  490 A provides an inductor  86 J  having inductance L to the L-C-L T-shaped resonator module  484 N. The AWM  490 A is then effectively coupled to an inductor  86 K with inductance -L forming the modified AWM  494 A (balanced inductance as shown by module  486 A in  FIG.  30 S ). 
       FIG.  30 U  is a simplified diagram of another embodiment  480 O that includes an IMM  484 N. Similar to architecture  480 F, architecture  480 O includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 N, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 N may include an acoustic wave module (AWM)  490 A coupled in parallel with a variable capacitor  98 A, and a post-impedance match component module (post-IMCM)  492 D. The variable capacitor  98 A may modify the capacitance of Cr and may be used to modify the impedance or resonate or anti-resonate of the AWM  490 A and thus the IMM  484 N. 
       FIG.  30 V  is a simplified diagram of another embodiment  480 P that includes an IMM  484 O. Similar to architecture  480 F, architecture  480 P includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 N, and an output load or impedance modeled by resistor  94 B. In an embodiment, the IMM  484 N may include an acoustic wave module (AWM)  490 A and a post-impedance match component module (post-IMCM)  492 D. The IMM  484 O may be configured to add impedance to architecture  480 P when output impedance is greater than the input impedance. In IMM  484 O, the post-IMCM  492 D may include a variable capacitor  98 C. The variable capacitor  98 C is coupled to ground and between the AWM  490 A and the output load  94 B. By varying the capacitor  98 C capacitance, the borrowed inductance from the AWM  490 A may also vary. Similarly the resonant frequency f r&#39;  and impedance may also vary. The variable capacitor  98 C may be used to tune the IMM  484 O resonant frequency and impedance in an embodiment. 
       FIG.  30 W  is a simplified diagram of another embodiment  480 Q that includes an IMM  484 P. Similar to architecture  480 N, architecture  480 O includes a signal source  92 A, an input load or impedance modeled by resistor  94 A, an IMM  484 P, and an output load or impedance modeled by resistor  94 B. As noted, the input load and output load  94 A,  94 B may include real and imaginary components. In an embodiment, the IMM  484 M may include a first acoustic wave module (AWM)  490 A coupled between the loads  94 A,  94 B and a second acoustic wave module (AWM)  490 B coupled between the first AWM  490 A and load  94 B at one end and ground at the other. 
     The AWM  490 A and AWM  490 B may be single acoustic wave devices or a plurality of devices and variable capacitors in various configurations as shown and described above. In an embodiment, the IMM  484 P may be configured to provide a balancing impedance to architecture  480 N at a desired or target frequency. The IMM  484 P may be configured to add impedance to architecture  480 Q when output impedance is not equal to the input impedance. As noted, input and output impedance  94 A,  94 B may include a real and imaginary imbalance. As shown in  FIG.  30 W , the first AWM  490 A may provide an effective inductor  86 J with inductance L. The inductance L may balance a real load difference between the input and output loads  94 A,  94 B and may balance an imaginary load differential.The borrowed inductance L via inductor  86 J may shift the effective inductance of the AWM  490 A by - L ( 86 K) as represented by the block  486 A. 
     Similarly, the second AWM  490 B may provide an effective capacitor  82 J with capacitance C. The capacitance C may balance a real load difference between the input and output loads  94 A,  94 B and may balance an imaginary load differential. The borrowed capacitance C from capacitor  82 J may shift the effective capacitance of the AWM  490 A by -C ( 82 K) as represented by the block  487 A. In an embodiment, the AWM  490 A and AWM  490 B may be selected to have a desired frequency response based on the borrowed or shifted inductance (AWM  490 A) or capacitance (AWM  490 B), the effect on a resonant or anti-resonant frequency of the AWM  490 A,  490 B due to the borrowed inductance or capacitance. For example, the AWM  490 A be configured to have a nominal frequency f r  that is shifted to resonate at f r&#39;  by the inductor  86 J. 
     Applications that may include the novel apparatus and systems of various embodiments include electronic circuitry used in high-speed computers, communication and signal processing circuitry, modems, single or multi-processor modules, single or multiple embedded processors, data switches, and application-specific modules, including multilayer, multi-chip modules. Such apparatus and systems may further be included as sub-components within a variety of electronic systems, such as televisions, cellular telephones, personal computers (e.g., laptop computers, desktop computers, handheld computers, tablet computers, etc.), workstations, radios, video players, audio players (e.g., mp3 players), vehicles, medical devices (e.g., heart monitor, blood pressure monitor, etc.) and others. Some embodiments may include a number of methods. 
     It may be possible to execute the activities described herein in an order other than the order described. Various activities described with respect to the methods identified herein can be executed in repetitive, serial, or parallel fashion. 
     A software program may be launched from a computer-readable medium in a computer-based system to execute functions defined in the software program. Various programming languages may be employed to create software programs designed to implement and perform the methods disclosed herein. The programs may be structured in an object-orientated format using an object-oriented language such as Java or C++. Alternatively, the programs may be structured in a procedure-orientated format using a procedural language, such as assembly or C. The software components may communicate using a number of mechanisms well known to those skilled in the art, such as application program interfaces or interprocess communication techniques, including remote procedure calls. The teachings of various embodiments are not limited to any particular programming language or environment. 
     The accompanying drawings that form a part hereof show, by way of illustration and not of limitation, specific embodiments in which the subject matter may be practiced. The embodiments illustrated are described in sufficient detail to enable those skilled in the art to practice the teachings disclosed herein. Other embodiments may be utilized and derived there-from, such that structural and logical substitutions and changes may be made without departing from the scope of this disclosure. This Detailed Description, therefore, is not to be taken in a limiting sense, and the scope of various embodiments is defined only by the appended claims, along with the full range of equivalents to which such claims are entitled. 
     Such embodiments of the inventive subject matter may be referred to herein individually or collectively by the term “invention” merely for convenience and without intending to voluntarily limit the scope of this application to any single invention or inventive concept, if more than one is in fact disclosed. Thus, although specific embodiments have been illustrated and described herein, any arrangement calculated to achieve the same purpose may be substituted for the specific embodiments shown. This disclosure is intended to cover any and all adaptations or variations of various embodiments. Combinations of the above embodiments, and other embodiments not specifically described herein, will be apparent to those of skill in the art upon reviewing the above description. 
     The Abstract of the Disclosure is provided to comply with 37 C.F.R. § 1.72(b), requiring an abstract that will allow the reader to quickly ascertain the nature of the technical disclosure. It is submitted with the understanding that it will not be used to interpret or limit the scope or meaning of the claims. In the foregoing Detailed Description, various features are grouped together in a single embodiment for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted to require more features than are expressly recited in each claim. Rather, inventive subject matter may be found in less than all features of a single disclosed embodiment. Thus the following claims are hereby incorporated into the Detailed Description, with each claim standing on its own as a separate embodiment.