Patent Publication Number: US-6339298-B1

Title: Dimming ballast resonant feedback circuit

Description:
FIELD OF INVENTION 
     The present invention relates to a ballast, or power supply circuit, for gas discharge lamps of the type using regenerative gate-drive circuitry to control a pair of serially connected, complementary conduction-type switches of a d.c.-a.c. inverter. More particularly, the invention relates to a resonant feedback circuit drawing continuous input current to satisfy requirements of phase control dimmers. 
     BACKGROUND OF THE INVENTION 
     Phase-controlled dimmable ballasts have gained a growing popularity in industry due to their capability for use with photo cells, motion detectors and standard wall dimmers. 
     Dimming of fluorescent lamps with class D converters is accomplished by either regulating the lamp current, or regulating the average current feeding the inverter. For cold cathode fluorescent lamps (CCFLs), the pulse width modulating (PWM) technique is commonly used to expand a dimming range. The technique pulses the CCFLs at full rated lamp current thereby modulating intensity by varying the percentage of time the lamp is operating at full-rated current. Such a system can operate with a closed loop or an open loop system The technique is simple, low cost, and a fixed frequency operation, however, it is not easily adapted to hot cathode fluorescent lamps. For proper dimming of hot cathode lamps, the cathode heating needs to be increased, as light intensity is reduced. If inadequate heating exists, cathode sputtering increases as the lamp is dimmed. Also, the lamp arc crest factor should be less than 1.7 for most dimming ranges, in order to maintain the rated lamp life. The higher the crest factor, the shorter will be the life of the lamp. The PWM method does not address these problems, and therefore so far has been limited to CCFL applications. 
     Class D inverter topology with variable frequency dimming has been widely accepted by the lighting industry for use in the preheat, ignition and dimming of a lamp. The benefits of such a topology include, but are not limited to (i) ease of implementing programmable starting sequences which extend lamp life; (ii) simplification of lamp network design; (iii) low cost to increase lamp cathode heating as the lamp is dimmed; (iv) obtainable low lamp arc crest factor; (v) ease of regulating the lamp power by either regulating the lamp current or the average current feeding the inverter; and (vi) zero voltage switching can be maintained by operating the switching frequency above the resonant frequency of the inverter. 
     Conventional class D circuits which are used for d.c.-to-d.c. converters or electronic ballasts, implement a two-pole active switch via two, n-channel devices or n-p-channel complementary pairs. A gate is voltage controllable from a control-integrated circuit (IC), which is normally referenced to ground, thus, the control signals have to be level shifted to the source of the high-side power device, which, in class D applications, swings between two rails of the circuit. The techniques presently used to perform this function are by either, transformer coupling or a high-voltage integrated circuit (HVIC) with a boot-strapped, high side driver. Either solution imposes a severe cost and performance penalty. 
     For transformer coupling, the transformer needs to have at least three isolated windings wound on a single core, adding to cost and space considerations. The windings need to be properly isolated to prevent breakdown due to the presence of high potential. Also, the gate&#39;s drive circuit needs to be damped and clamped to prevent ringing between leakage inductors of the transformer and parasitic capacitors of switching MOSFETs. 
     In the case of high-voltage integrated circuits (HVIC), the HVIC has two isolated output buffers and logic circuitry which is sensitive to negative transients. The high-voltage process for the IC increases the size of the silicon die, and the boot-strap components add to the part count and costs. Such a system is also severely limited as to the switching frequency obtainable, which commonly is less than 100K Hz. Consequently, it uses the large sizes of EMI filters and resonant components and requires larger space for implementation. 
     In incandescent lamp dimming systems, dimming is controlled by a phase dimmer, also known as a triac dimmer. A common type of phase dimmer, blocks a portion of each positive or negative half cycle immediately after the zero crossing of the voltage. The clipped waveform carries both the power and dimming signal to the loads. The dimmer replaces a wall switch which is installed in series with a power line. 
     It would be desirable to use existing phase dimmer signals for dimming of compact fluorescent lamps (CFL). A system designed to use existing triac phase dimmers must satisfy the requirements of the triac, one of which is a holding current specification. When the triac is in a conducting state, the current through the triac must remain above the specified holding current in order for the triac not to switch off and interrupt current. It would also be desirable to have such a system use a single-stage design for dimming and interfacing with a phase dimmer, provided at a low cost, with a direct gate drive for both high and low side MOSFET switches, with minimal voltage and current stresses on a resonant circuit. Still a further desirable aspect is to have a circuit which would allow programmable starting sequences to extend a lamp life, allow for low lamp arc crest factors and zero voltage switching over wide ranges. Such a system should also include compact size with low component counts and be easily adapted for different line input voltage and powers and provide for adequate protection for abnormal operations. 
     SUMMARY OF THE INVENTION 
     In an embodiment of the present invention, a dimmable ballast circuit is designed to receive a phase dimmer signal to control output of a fluorescent lamp. The dimming ballast includes an input section configured to receive the phase dimmer signal. The system includes a low cost integrated chip having an internal operational amplifier with a non-inverting input tied to a steady-state input within the integrated chip for a totem pole output. The IC is also configured in a floating ground arrangement with the floating ground connected to the inverting input of the operational amplifier. A coupling capacitor is connected at one end of the output of the controller IC. A switching network is designed with a pair of complementary connected switches, and is also connected to receive the output from the IC through a second end of the coupling capacitor. A current-sensing resistor is used to sense the switching current of a power switch in order to generate a feedback signal. A level shifter is designed to receive a signal from the input section, and to shift the received signal from a level of the reference ground to a level of the floating ground, the error difference between level shifted signal and the feedback signal are amplified by a separate operational amplifier not part of the IC, and the amplified signal is supplied to the frequency control input of the integrated chip. In this manner the output frequency of the integrated chip regulates the output intensity of the lamp. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a simplified schematic illustrating the concept of resonant feedback; 
     FIG. 2 is an improved version of the schematic depicted in FIG. 1; and 
     FIG. 3 is a detailed schematic of one embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 shows a partial schematic of a ballast dimmer circuit employing a first embodiment of the present invention. Shown in FIG. 1 are: a phase dimmer input or source  10 , an input network  12 , a resonant circuit load  14 , complementary conduction-type switches  16  and resonant feedback circuit  18 . Omitted from FIG. 1 for the sake of simplicity are an EMI filter ( 220  in FIG.  3 ), a level shifting circuit ( 60  in FIG.  3 ), a compensation network ( 62  in FIG.  3 ), a controller integrated circuit ( 64  in FIG. 3) and a load sensing circuit ( 112  in FIG.  3 ). 
     Positive going signals from phase dimmer source  10  are rectified through first and second diodes  20  and  22  connected in series with capacitor  24  whose remaining lead is connected to the remaining phase dimmer source  10 . Negative going signals from phase dimmer source  10  are rectified through third and fourth diodes  26  and  28  connected in series with capacitor  30  whose remaining lead is connected to the remaining phase dimmer source  10  input lead which places capacitors  24  and  30  in a series voltage-doubler arrangement. Complementary FET switches  32  and  34  are connected in a common source arrangement across voltage-doubling capacitors  24  and  30 . The gates of switches  32  and  34  are connected to common drive signal  35  that is, in turn, connected to common source node  36 . Drive signal  35  is a variable frequency, square wave drive signal, alternately positive and then negative with respect to common source node  36 . 
     Resonant inductor  37  is connected to the common source leads of switches  32  and  34  and to node  38  which is a common connection node of resonant capacitors  40  and  42 . Resonant capacitor  40 , is in turn connected to the junction between diodes  20  and  22  and second resonant feedback capacitor  42  is connected at the junction of diodes  26  and  28 . Capacitor  44  is connected between the two aforementioned diode junctions so that it is bridging capacitors  40  and  42 . Resonant load capacitor  46  is connected between node  38  and the anode lead of diode  28  which also serves as ground node  48 . The load being powered by resonant load circuit  14  comprises capacitor  50  connected in series with lamp  52  between nodes  38  and  48 . 
     The described circuit is especially beneficial for an electronic ballast or phase dimmer circuit, such as a triac dimmer, employing an EMI filter. It is known that if an EMI filter included with the electronic ballast or the phase dimmer is not properly loaded, there is a danger it could misfire the dimmer causing the lamp to flicker. The just described circuit acts to continue loading the EMI circuit and eliminate lamp flicker even at low conduction angles. 
     Since the purpose of this discussion is to explain the functioning of resonant feedback circuit  18 , explanation of a complete ballast dimming circuit will be deferred until later when FIG. 3 is explained. For the purposes of this discussion it can be assumed, as described above, that square wave signal  35  is connected between common source node  36  and the common gate connections of switches  32  and  34  thus causing common source node  36  to be alternately switched between ground potential at node  48  and the full DC potential at the cathode of diode  22 . Under steady-state operation, it can also be assumed that capacitors  24  and  30  have acquired a fall working charge and that resonant load circuit  14  has stored energy in resonant inductor  37  and resonant capacitor  46 . Under these conditions, because the switching frequency at node  36  is many times the frequency of phase dimmer source  10 , without a feedback circuit or other special circuits, current load on the phase dimmer would fall below the minimum holding current for the phase dimmer triac causing the triac to switch off prematurely. Resonant feedback capacitors  40  and  42  provide an economical design for exchanging energy between resonant load circuit  14  and input network  12  so that current drawn from phase dimmer source  10  does not fall below its minimum holding current at any time within the conduction phase of the dimmer&#39;s triac. 
     While the circuit of FIG. 1 is effective in meeting minimum input holding current requirements, still further improvements are possible in terms of minimizing the lamp current crest factor. To understand how an improvement in crest factor is possible, it is necessary to understand the effect of feedback capacitors  40  and  42  on the resonant load circuit  14 . This can best be accomplished by presenting an equivalent input capacitance approximation given by 
     
       
           C   eq =( C   40   +C   42 )(| v   in |+2 V   a   −V   dc1 )/2 V   a   (1) 
       
     
     where v in  is the phase dimmer source  10  input voltage, V a  is the peak AC voltage on resonant capacitor  46 , v dc1  is the voltage on capacitor  24 , C 40  is capacitor  40  and C 42  is capacitor  42 . It can be seen from Equation (1) that C eq  is highly dependent on the magnitude of the phase dimmer input voltage, and, due to the effect of C eq , the total resonant capacitance (C eq +capacitor  46 ) varies with the input voltage. As a result, the crest factor of the lamp current can be higher than desirable. The chopped nature of the line waveform from a phase dimmer makes the problem even more pronounced. 
     FIG. 2 shows a second embodiment of the present invention with an improved circuit in terms of lamp current crest factor. The second embodiment is similar to that in FIG. 1, and like numbered components in FIG. 2 serve the same purpose in both figures. Resonant feedback capacitor  40  in FIG. 1 has been removed in FIG. 2, and resonant feedback capacitor  56  has been added in FIG. 2, in parallel with diode  22 . By properly selecting C 56  and C 42 , the variation of C eq  will be minimized, thus minimizing the effects of resonant feedback capacitors C 56  and C 42  on the resonant tank  14  and on the lamp crest factor. 
     Turning now to FIG. 3, illustrated is a more complete schematic incorporating the improvements shown in FIG. 2 into a floating IC driven ballast. Like numbered numerals in FIGS. 2 and 3 identify components serving identical purposes. Since like numbered components in FIG. 3 function exactly as described for FIG. 2, their function will not be described again in the following discussion. Similarly, since the finctioning of level shifting circuits, compensation networks and controller IC circuits like level shifting circuit  60 , compensation network  62  and controller IC circuit  64  are well understood in the art, they will not be described in detail here. Their function will, however, be described sufficiently to understand their interaction with the concepts of the present invention. 
     Input phase dimmer voltage source  10  generates a bus voltage  66 , and a phase dimmer signal  68 . Node  48  serves as ground reference for the ballast circuit. Bus voltage  66  is provided to a switching network  70 , and phase dimmer signal  68  is provided to level shifting circuit  60  having a floating ground reference comprising common source node  36 . A controller integrated circuit (IC)  72 , such as a current mode pulse width modulated (PWM) controller IC, delivers a gate drive  74  to switches  32  and  34  through the coupling capacitor  76 . In the present embodiment switches  32 ,  34  may be configured as a complementary pair of MOSFETs, with switch  32  being an n-channel MOSFET and switch  34  being a p-channel MOSFET. Controller IC  72  is configured with a floating ground  78 , corresponding to node  36 , and is supplied with a compensation network  62 , and IC  72  supplies a reference voltage  80 . The IC  72  is powered by a signal from a voltage source  82 . Phase dimmer signal  68  is therefore a chopped input voltage which is shifted from circuit ground to a floating signal ground. 
     Switching network  70  delivers signals to a load circuit  84  having a series resonant configuration including resonant inductor  37  in series with resonant capacitor  46 . A matching capacitor  86  is provided for low bus applications in order to maintain sufficient voltage as lamp  88  is dimmed, with the lamp cathodes heating being powered through windings  90  and  92 . Lamp  88  may, in one embodiment, be a compact fluorescent lamp. 
     Resistors  94  and  96  work in conjunction with voltage source  82  in order to ensure proper start-up of controller IC  72 . The parallel combinations of diode  98 , resistor  100  and diode  102 , resistor  104  provides sufficient dead time to complementary switches  32  and  34 , respectively. Resistor  105  works in conjunction with capacitor  76  to convert the pulse DC output of the IC  72  to an AC square waveform through diode  98 , resistors  100 ,  104 , and diode  102  in order to drive the switches  32  and  34 . Resistor  105  is important because it provides the initial charging of capacitor  72 , and, therefore, determines the initial time delay until a transition to normal switching of switches  32  and  34  occurs. When the circuit is first activated, only switch  32  will be biased to the on state because the output on pin  6  of integrated circuit  72  is always positive with respect to floating ground at nodes  78  and  36 . As capacitor  72  charges, the current through resistor  105  will transition gradually from a current substantially in one direction to an alternating square wave current. At this time switches  32  and  34  will be alternately switched on and off. This transition must occur before capacitor  106  loses much of its initial charge because capacitor  106  is the initial source of energy for powering integrated circuit  72 . If, for example, capacitor  106  is initially charged to 16 volts, transition to normal switching of switches  32  and  34  must occur before the charge on capacitor  106  falls below 9 volts. Additional details on the source of power for integrated circuit  72  are discussed later. 
     The network of capacitor  107  and resistor  108  function as a low pass filter to provide an average current feedback signal  110 , based on the output of current sense resistor  112 , so as to provide current feedback signal  110  to compensation network  62 . Current sense resistor  112  has parallel diodes  116  and  118  connected across it in opposite directions to limit the voltage drop across it to not more than 0.7 volts. In this way switches  32  and  34  are always operated in the saturation region and are protected from operating in the linear region during startup which can cause overheating and failure of the switches. 
     Switching network  70  has a common to ground  48 , and the point between switch  32  and switch  34  nearest switch  34  is at floating ground  36 . 
     Whereas the potential of circuit ground such as circuit grounds  48  and  120  are unchanging, the potential of a floating ground, such as that comprising nodes  36 ,  78 ,  122 ,  124 ,  126  and  128 , are constantly changing with reference to the circuit grounds. Thus, when switch  34  is turned on, floating ground  36  will be moved to circuit ground. However, when switch  32  is turned on, floating ground  36  will become substantially equivalent to the bus voltage value  66 . Further, since the floating ground nodes are tied together, controller IC  72  also varies between these levels. 
     Use of the floating ground configuration allows the use of a low voltage IC, such as a 35-volt IC instead of a more expensive high-voltage IC. Also, by implementing the low-voltage IC, a transformer coupling the gate drives is not necessary. Further, using the floating ground IC technique, it is possible to drive the ballast circuit into the megahertz range since power dissipation on the IC is extremely low compared to high-voltage techniques. 
     A challenge faced when implementing the present design of using a floating ground reference for controller IC  72 , is a manner of desirably delivering dimming signal  68  to controller IC  72 . This is a challenge since the floating ground value swings from ground reference to substantially the bus voltage input. In the present invention, dimming signal  68  is provided to controller IC  72  through level shifter circuit  60 , which is provided with a floating ground  126 , tied to the floating ground  78  of controller IC  72 . By this arrangement, a signal provided from the rectified input dimmer voltage source  10 , which is tied to circuit ground  120 , may—as shown in level shifter circuit  60 —be shifted through Zener diode  130 , resistors  132 ,  134 ,  136  and Zener diode  140 . Capacitors  144  and  146  and resister  148  also comprise a portion of the level shifting circuit. Diode  150  is connected between zener diode  140  and capacitor  146  at one end, and to operational amplifier  152  at its other end. 
     The present invention further uses a current sensing technique to provide the desired output under the constraints of controller IC  72 . In particular, current sensing resistor  112  is used to obtain actual lamp system power. Capacitor  107  and resistor  108  provide the average value of the switching current when the bus voltage is fixed. Using an average value of the bus voltage times the average value of switching current, the system power can be controlled and therefore also, the lamp lumen output. It is noted that the average current of the system is that detected through resistor  108 , and obtaining the average value of the bus voltage may be achieved by various known techniques. By lowering system power, light output of lamp  88  will be lowered and by increasing system power light output of lamp  88  is increased. 
     Using the floating ground system configuration of the present embodiment means feedback signal  110  will be a positive signal. Positive feedback signal  110  is fed to the inverting input of operational amplifier  152  of compensation network  62 . Compensation network  62  further comprises resistors  154  and  156 , capacitor  158 . The non-inverting input of operational amplifier  152  receives its input through resistor  148  of level shifting circuit  60 . The output of operational amplifier  152  is then provided to controller IC circuit  64  at the base terminal of transistor  162  which in turn varies the effective resistance of the timing resistor connected between pins  8  and  4  of the controller IC  72 . Controller IC circuit  64  further includes transistor  164 , resistors  166 ,  168 ,  170 , capacitors  172 ,  174 ,  176 ,  178 ,  180 , diodes  182 ,  184 , Zener diode  186  and controller IC  72 . Operation of this circuit acts to adjust the output frequency of controller IC  72  at pin  6  to coupling capacitor  76  and through resistor  105  to floating ground  36 , and thereby maintain the lumen output at a given dimming level. 
     The present invention uses a complimentary pair of MOSFETs driven by controller IC  72  through a.c.-coupling capacitor  76  to operate lamp  88 . The driving scheme eliminates the need for a high-side driver or a pulse transformer and/or generating a negative bias gate or other driving scheme. 
     A further mentioned concept of the present invention is the use of level-shifting circuit  60  which shifts chopped dimming signal  68 , from a ground reference level of voltage source  10  to a floating ground signal. The shifting of this dimming signal  68  allows the input signal from level shifter  60  to be used by controller IC  76 . 
     With attention to input section  12 , phase dimmer source  10  is connected to supply resistive inductive components  200  and  202 , respectively. An RC network comprised of capacitor  204  and resistor element  206  are placed across the inputs of the voltage doubling rectifier circuit which is comprised of diodes  20 ,  22 ,  26 ,  28  and capacitors  24  and  30 . Capacitor  204  and resistor element  206  cooperate with inductor  202  to form an EMI filter  220 . The rectified phase dimmer signal  68  is supplied to level shifter circuit  60  via Zener diode  130 . Zener diode  130  is supplied to ensure appropriate voltage levels, especially in light of the voltage doubling rectifier circuit configuration. 
     Turning attention to the voltage source  82  which supplies voltage to controller IC  72  on pin  7 , a network comprising resistors  222 ,  224  and  226 , diode  228 , capacitors  230 ,  106  and  234 , and inductor  238  form a start-up circuit  240 , to generate the necessary voltage for starting of controller IC  72 . It is noted that once controller IC  72  is charged up to an operating voltage, controller IC  72  will consume more power than can be supplied by the described start-up circuit  240  through resistors  222  and  224 . Therefore, further DC bias is provided by inductor  238 . Inductors  37 ,  90 ,  92  and  238  all share the same core, poled as indicated by dots on the schematic in FIG.  3 . 
     The above-described circuit provides a voltage-fed series resonant class D system with variable frequency, which is particularly applicable for use in compact fluorescent lamps. This topology allows easily operating in zero-voltage switching (ZVS) resonant mode, reduces the MOSFET switching losses and electrical magnetic interference. Further, by varying the switching frequency, it is possible to modulate the average current in the switching MOSFETs and therefore the output power. 
     The complementary pair of MOSFETs  32 ,  34  of the present embodiment are driven by a low-cost, single totem pole, class D, buffer output, such as a UC3844A or equivalent controller IC  72 , through a.c. coupling capacitor  76 . The cascade class D driving scheme eliminates the need for a high-voltage integrated chip (HVIC) or a pulse transformer and/or generating a negative gate bias. The technique is capable of providing switching frequency up to the megahertz range. Appropriate fusing elements are also depicted in FIG.  3 . 
     Exemplary component values and/or designations for the circuit of FIG. 3 are as follows for a compact fluorescent lamp rated at 28 watts with a d.c. bus voltage of at least 120 volts: 
     
       
         
           
               
               
               
               
             
               
                   
                   
               
             
            
               
                   
                 Capacitors 24, 30 
                 22 
                 micro-farads 
               
               
                   
                 Inductor 37 
                 1 
                 milli-henry 
               
               
                   
                 Capacitor 42 
                 1500 
                 pico-farads 
               
               
                   
                 Capacitor 56 
                 0.0022 
                 micro-farads 
               
               
                   
                 Capacitor 44 
                 0.1 
                 micro-farads 
               
               
                   
                 Capacitor 46 
                 1500 
                 pico-farads 
               
               
                   
                 Capacitor 86 
                 0.1 
                 micro-farads 
               
               
                   
                 Resistors 94, 96 
                 75K 
                 ohms 
               
               
                   
                 Resistors 100, 104 
                 700 
                 ohms 
               
               
                   
                 Capacitor 107 
                 .001 
                 micro-farads 
               
               
                   
                 Resistor 108 
                 3.3K 
                 ohms 
               
               
                   
                 Resistor 112 
                 5.1 
                 ohms 
               
               
                   
                 Resistor 132 
                 1.5M 
                 ohms 
               
               
                   
                 Resistor 134 
                 200K 
                 ohms 
               
               
                   
                 Resistor 136 
                 30K 
                 ohms 
               
               
                   
                 Resistor 138 
                 10K 
                 ohms 
               
               
                   
                 Capacitor 144 
                 0.1 
                 micro-farads 
               
               
                   
                 Capacitor 146 
                 0.22 
                 micro-farads 
               
               
                   
                 Resistor 148 
                 3M 
                 ohms 
               
               
                   
                 Resistor 154 
                 300K 
                 ohms 
               
               
                   
                 Resistor 156 
                 240K 
                 ohms 
               
               
                   
                 Capacitor 158 
                 0.01 
                 micro-farads 
               
               
                   
                 Resistor 166 
                 150K 
                 ohms 
               
               
                   
                 Resistor 168 
                 7.5K 
                 ohms 
               
               
                   
                 Resistor 170 
                 8K 
                 ohms 
               
               
                   
                 Resistor 172 
                 1K 
                 ohms 
               
               
                   
                 Capacitor 174 
                 0.001 
                 micro-farads 
               
               
                   
                 Capacitors 176, 178 
                 390 
                 pico-farads 
               
               
                   
                 Capacitor 180 
                 10 
                 micro-farads 
               
               
                   
                 Resistor 105 
                 20K 
                 ohms 
               
               
                   
                 Resistor 200 
                 5.1 
                 ohms 
               
               
                   
                 Inductor 202 
                 2.5 
                 milli-henries 
               
               
                   
                 Capacitor 204 
                 0.1 
                 micro-farads 
               
               
                   
                 Resistor 206 
                 330 
                 ohms 
               
               
                   
                 Resistors 222, 224 
                 75K 
                 ohms 
               
               
                   
                 Resistor 226 
                 100 
                 ohms 
               
               
                   
                 Capacitor 228 
                 0.1 
                 micro-farads 
               
               
                   
                 Capacitor 230 
                 22 
                 micro-farads 
               
               
                   
                 Capacitor 106 
                 0.1 
                 micro-farads 
               
               
                   
                   
               
            
           
         
       
     
     In addition, MOSFET  32  is sold under the designation IRF310, MOSFET  34  under designation IRF9310, transistors  162  and  164  under designation FMB3946. Diodes  98 ,  102 ,  116 ,  118 ,  150 ,  182 , and  184  are sold under designation 1N4148, and diodes  20 ,  22 ,  26  and  28  under designation RGL41J. 
     While the invention has been described with respect to specific embodiments by way of illustration, many modifications and changes will occur to those skilled in the art. It is therefore, to be understood that the appended claims are intended to cover all such modifications and changes which fall within the true spirit and scope of the invention.