Patent Publication Number: US-2009219002-A1

Title: Switching power supply device and a semiconductor integrated circuit

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This is a continuation of U.S. Ser. No. 11/979,093, filed Oct. 31, 2007, which is a continuation application of U.S. Ser. No. 11/510,819, filed Aug. 28, 2006 (now U.S. Pat. No. 7,307,406). 
    
    
     The present application claims priority from Japanese patent application No. 2005-248317 filed on Aug. 29, 2005 the content of which is hereby incorporated by reference into this application. 
     BACKGROUND OF THE INVENTION 
     The present invention relates to a switching power supply device and a semiconductor integrated circuit, more particularly to, for instance, a technique that can be effectively applied to a switching power supply device for converting a high voltage into a low voltage. 
     Examples of transformer type synchronous rectifying converter include what are disclosed in the Japanese Unexamined Patent Publications Nos. 2001-346380 and 2001-008444. 
     [Patent Reference 1] Japanese Unexamined Patent Publication No. 2001-346380 
     [Patent Reference 2] Japanese Unexamined Patent Publication No. 2001-008444 
     SUMMARY OF THE INVENTION 
     Switching power supply devices are required to be inexpensive, compact and efficient, operate on a low voltage and provide a large current. For this reason, they often use as switch elements N-channel type power MOSFETS (hereinafter abbreviated to NMOSs), which are inexpensive, low in on-resistance (low Ron) and in the quantity of gate charge (low Qgd).  FIG. 7  shows a block diagram of a voltage step-down type switching power supply device studied before the invention of the present application. The switching power supply device shown in  FIG. 7  supplies a current to the input side of an inductor L 1  via a high potential side switch MOSFET Q 1  which is subjected to switch control with a pulse width modulation (PWM) signal and, provided with an output capacitor Co between the output side of the inductor L 1  and the ground potential of the circuit, obtains an output voltage Vout. Between the inductor L 1  and the ground potential, a low potential side switch MOSFET Q 2  is provided. This MOSFET Q 2  causes the input side of the inductor L 1  when the MOSFET Q 1  is turned off to be voltage-clamped to the ground potential of the circuit. The MOSFETs Q 1  and Q 2  are alternately, and their midpoint voltage Vsw manifests a waveform reciprocating between 0 V and an input voltage Vin. Stabilization of the output voltage Vout is achieved by adjusting the duty of PWM. More specifically, a PWM controller (not shown) is used to generate a PWM signal matching the output voltage Vout, and that signal is given to a driver DVIC. 
     A continuous current mode (under heavy load) and a reverse current mode (under light load) in the voltage step-down type switching power supply device will be described.  FIG. 8  shows the switching waveform in the continuous current mode and  FIG. 9  shows the switching waveform in the reverse current mode. In the case of the continuous current mode shown in  FIG. 8 , a current IL flowing to the inductor (choke coil) L 1  always constitutes a triangular wave of a positive value in at least one PWM cycle (a PWM signal is used in this case, though it need not be a PWM signal but any signal that controls the output voltage Vout by controlling the switching of power MOSFETs, such as a pulse frequency modulation (PFM) signal or a pulse density modulation (PDM) signal can be used), and its average is equal to the output current Iout. When the output current Iout becomes smaller, the current IL drops on the whole. And it is seen that there are periods of negative values (in the reverse direction to the current I 2  in the same graph as indicated by solid black parts in  FIG. 9 . These are periods when a current is flowing in the reverse direction from the output capacitor Co to the MOSFET Q 2  via the inductor L 1 . 
     From the turn-off of the high potential side MOSFET TQ 1  until that of the low potential side MOSFET Q 2  and from the turn-off of the MOSFET Q 2  until that of the MOSFET Q 1 , periods in which both are turned off are set to prevent a through current from flowing by the simultaneous turning-on of both MOSFETS. Such a period is generally known as a dead time. Since both MOSFETs Q 1  and Q 2  are off during this dead time, the output current Iout during the period flows to the load side via the body diode (parasitic diode between the source and the substrate) of the MOSFET Q 2 . As the equivalent resistance of the body diode is higher than the on-resistance of the MOSFET Q 2 , usually the dead time is designed to be as short as practicable with a view to higher circuit efficiency, and its length is constant whether in the continuous current mode or in the reverse current mode. The inventors of the present application intend to improve efficiency by a contrivance oriented to this reverse current mode. 
     An object of the present invention is to provide a switching power supply device and a semiconductor integrated circuit realizing such efficiency improvement. This and other objects and novel features of the invention will become apparent from the description in this specification when taken in conjunction with the accompanying drawings. 
     A typical one of the aspects of the invention disclosed in this application will be briefly summarized below. A capacitor is disposed between the output side and the ground potential of an inductor which creates an output voltage. A first switch element supplies a current from an input voltage to the input side of the inductor, and a second switch element which comes on when the first switch element is off sets the input side of the inductor to a prescribed potential A control circuit detects the arrival of the voltage on the input side of the inductor at a high voltage corresponding to the input voltage when a load circuit is in a light load state and the second switch element is turned off, and turns on the first switch element. When the load circuit is in a heavy load state, it invalidates the detection output of the voltage detecting circuit and, after the second switch element is turned off, turns on the first switch element. 
     The reverse current in a light load state can be utilized for charging the parasitic capacitance on the input side of the inductor, and the turn-on loss at the first switch element can be substantially reduced. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic circuit diagram showing a switching power supply device, which is a preferred embodiment of the present invention. 
         FIG. 2  shows switching waveforms in the switching power supply device of  FIG. 1  when in the reverse current mode. 
         FIG. 3  illustrates loss analysis in switching power supply devices according to the related art and the present invention under light load. 
         FIG. 4  illustrates circuit efficiency in the switching power supply devices according to the related art and the invention. 
         FIG. 5  is a circuit diagram showing a switching power supply device, which is another preferred embodiment of the invention. 
         FIG. 6  is a schematic overall circuit diagram showing a switching power supply device, which is still another preferred embodiment of the invention. 
         FIG. 7  is a block diagram of a step-down type switching power supply device studied prior to the invention of the present application. 
         FIG. 8  shows switching waveforms in the switching power supply device of  FIG. 7  when in the continuous current mode. 
         FIG. 9  shows switching waveforms in the switching power supply device of  FIG. 7  when in the reverse current mode. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 1  is a schematic circuit diagram showing a switching power supply device, which is a preferred embodiment of the present invention. This embodiment is intended for a so-called step-down type switching power supply device which forms an output voltage Vout, stepped down from an input voltage Vin. The input voltage Vin is supposed to be a relatively high voltage, such as about 12 V, and the output voltage Vout, a relatively low voltage, such as about 1.3 V, though not particularly limited to these voltages. 
     For the input voltage Vin, a current  11  is supplied from the input side of an inductor L 1  via a high potential side switch MOSFET Q 1 . A capacitor Co is disposed between the output side of the inductor L 1  and the ground potential GND and the circuit, and the output voltage Vout smoothed by this capacitor Co is formed. This output voltage Vout serves as the operational voltage for a load circuit, such as a microprocessor of a CPU. A switch MOSFET Q 2  is provided between the input side of the inductor L 1  and the ground potential GND of the circuit. This MOSFET Q 2  comes on when the switch MOSFET Q 1  is off, brings the midpoint voltage Vsw to the ground potential of the circuit and clamps the counter electromotive voltage generated in the inductor L 1 . The switch MOSFETs Q 1  and Q 2  are composed of N-channel type power MOSFETs. As stated above, the connection point of the switch MOSFETs Q 1  and Q 2  is connected to the input side of the inductor L 1 . 
     Though not illustrated in  FIG. 1 , a PWM signal which is formed by a PWM generating circuit and controls the output voltage Vout to about 1.3V is entered into an input control circuit CONT. The input control circuit CONT forms a high voltage signal HC and a low potential side signal LC matching the PWM signal. Dead times are set for the two signals HC and LC. The high potential side signal HC is conveyed to a gate circuit G 1  through a level shift (level converting) circuit LS and to the gate of the high potential side switch MOSFET Q 1  through a driver DV 1 . The low potential side signal LC is conveyed to the gate of the high potential side switch MOSFET Q 2  through a driver DV 2 . 
     In this embodiment, a low Ron and low Qgd N-channel type power MOSFET Q 1  is used as the high potential side switch element, which is operated as a source follower output circuit. For this reason, a booster circuit is provided to obtain a high enough voltage as the midpoint potential to match the input voltage Vin, or in other words to prevent the midpoint potential Vsw from falling as much as the threshold voltage of the MOSFET Q 1  and there inviting a loss. 
     The booster circuit so operates as to boost the gate voltage to a level higher than the input voltage Vin by more than the threshold voltage of the MOSFET Q 1  when it is on. Thus, the midpoint is connected to one end of a boot strap capacitance CB. The other end of this boot strap capacitance CB is connected to a power supply terminal Vcc via a diode D 1 . The power voltage supplied from the power supply terminal Vcc is a low voltage, such as about 5 V, and is used as the operational voltage for the input control circuit CONT, the low potential side circuit of the level shifter LS, the driver DV 2  and a logic circuit LOG to be described afterwards. When the MOSFET Q 1  is off and the MOSFET Q 2  is on, the boot strap capacitance CB is charged up from the power supply terminal Vcc. When the MOSFET Q 2  is turned off and the MOSFET Q 1  is turned on, the gate voltage is boosted above the source side potential by the charged-up voltage for the boot strap capacitance CB. 
     This embodiment is provided with voltage dividing resistances R 1  and R 2  for dividing the input voltage. The resistance ratio between these voltage dividing resistances R 1  and R 2  is set to 1:4 or the like, though the ratio is not limited to this, and forms a divided voltage corresponding to 80% of the power voltage Vin. A voltage comparator circuit CMP compares the divided voltage and the midpoint voltage Vsw. When the midpoint voltage Vsw becomes higher than the divided voltage, the voltage comparator circuit CMP creates a detection signal and sends it to the logic circuit LOG. The logic circuit LOG, receiving a light load/heavy load mode signal MOD from a load circuit, controls the validity/invalidity of the detection signal of the voltage comparator circuit CMP, though its function is not limited to this. Thus, when the light load mode is indicated, the detection signal of the voltage comparator circuit CMP is validated. When the heavy load mode is indicated, the detection signal of the voltage comparator circuit CMP is invalidated. 
     In this embodiment, the input control circuit CONT, the level shift circuit LS, the gate circuit G 1 , the logic circuit LOG, the drivers DV 1  and DV 2 , the voltage comparator circuit CMP and the voltage dividing resistances R 1  and R 2  are formed over a single semiconductor substrate to serve as a control circuit DVIC. Therefore, a terminal T 1  to which the boot strap capacitance CB is connected, a terminal T 2  to which the input voltage Vin is inputted, a terminal T 3  to which the gate of the MOSFET Q 1  is connected, a terminal T 4  to which the gate of the MOSFET Q 2  is connected, a terminal T 5  to which the light load/heavy load mode signal MOD is inputted, a terminal T 6  to which the PWM signal is inputted and a terminal T 7  to which the power voltage Vcc is supplied are provided as external terminals. 
     Incidentally, a single semiconductor integrated circuit may as well be configured of the MOSFET Q 1  formed over a first semiconductor substrate, the MOSFET Q 2  formed over a second semiconductor substrate, the control circuit DVIC formed over a third semiconductor substrate, the control circuit DVIC, the MOSFET Q 1  and the MOSFET Q 2  being encapsulated in a single package. Alternatively, a single semiconductor integrated circuit may be configured of the control circuit DVIC and the MOSFET Q 1  together formed over the first semiconductor substrate and the MOSFET Q 2  formed over the second semiconductor substrate, the control circuit DVIC, the MOSFET Q 1  and the MOSFET Q 2  being encapsulated in a single package. 
       FIG. 2  shows switching waveforms when in the reverse current mode, in which in the switching power supply device according to the invention is under light load. The switching between the turning-off of the low potential side MOSFET Q 2  and the turning-on of the high potential side MOSFET Q 1  involve, as shown in the expanded part of  FIG. 2 , when the low potential side MOSFET Q 2  is turned off, there are periods of negative values as indicated by solid black parts as in the foregoing. If the high potential side MOSFET Q 1  is kept off then, the negative currents −12 will enable the parasitic capacitance between the midpoint the ground potential of the circuit to be utilized for charging and thereby to let the midpoint potential Vsw rise. 
     In the embodiment of  FIG. 1 , as the arrival of this midpoint potential Vsw at about 80% of the input voltage is detected by the voltage comparator circuit CMP and the high potential side MOSFET Q 1  is turned on through the logic circuit LOG-gate circuit G 1  and the driver DV 1 , the MOSFET Q 1  is turned on at a timing when the midpoint potential Vsw has become substantially equal to the input voltage Vin with the delay time on the signal path being taken into account. This makes it possible to reduce the power needed to let the midpoint voltage Vsw rise to the input voltage Vin to zero. In other words, the turn-on loss required to let the midpoint voltage Vsw from 0 V to the input voltage Vin when the MOSFET Q 1  is turned on can be represented by Equation (1) below. 
       Turn-on loss=½ ×Cx×V in 2   ×f  (where Cx is the parasitic capacitance between the midpoint and the ground potential GND of the circuit and f is the switching frequency) 
     As stated above, when in the reverse current mode, there are periods during which the current flows back from the output capacitor Co to the low potential side MOSFET Q 2 , and when the low potential side MOSFET Q 2  is turned off, that current charges parasitic capacitance between the drain and source of the low potential side MOSFET (between the midpoint and the ground potential GND of the circuit). In this embodiment, the midpoint voltage Vsw after the low potential side MOSFET Q 2  is turned of when in the reverse current mode is monitored by the voltage comparator circuit CMP, and when the midpoint voltage Vsw has substantially reached the input voltage Vin (for instance, a potential of 80% of Vi), the high potential side MOSFET Q 1  is turned on. 
     Depending on the level of the reverse current, the midpoint voltage Vsw may not reach the input voltage Vin. In such a case, as the voltage comparator circuit CMP forms no detection signal, a maximum limit is imposed on the length of the dead time between the turning-off of the low potential side MOSFET Q 2  until the turning-on off of the high potential side MOSFET Q 1 . In the embodiment of  FIG. 1 , the maximum dead time≦about 50 ns is provided in the logic circuit LOG (permissible time setting circuit), and after the lapse of this time the high potential side MOSFET Q 1  is turned on by way of the gate circuit G 1 -driver DV 1 . 
       FIG. 3  illustrates loss analysis in switching power supply devices according to the related art and the present invention under light load. The circuit conditions are supposed to be the input voltage Vin=12 V, the output voltage Vout=1.3 V, the output current Iout=1.0 A, the frequency f=500 KHz and the inductor L 1 =0.45H. There are eight types of losses including (1) Q 1  turn-off loss, (2) Q 1  turn-on loss, (3) body diode loss, (4) Q 2  conduction loss, (5) Q 1  conduction loss, (6) 02 drive loss, (7) Q 1  drive loss, (8) driver I C loss. Of these, (2) the Q 1  turn-on loss is as stated above, and other losses can be represented as follows. 
       Turn-off loss=0.5× V in×( I out+0.5 ×Ipp ) 2   ×Ins/A×f   (1) 
       Body diode loss= TD/TS×VF ×( I out+0.5 ×Ipp ) (where TD is the dead time, TS is the cycle and VF is the voltage of the body diode in the forward direction)  (3) 
       Conduction loss=( I out×Duty×√(1+⅓(0.5× Ipp/I out)) 2 ) 2   ×R on (where Ipp is the ripple current of IL, and Ron is the on-resistance of MOSFET)  (4), (5) 
       Drive loss= Qg×Vg×f  (where Qg is the gate charge of MOSFET and Vg is the gate drive voltage)  (6), (7) 
       Driver loss= Icc×Vcc  (where Icc is the self-consumed current and Vcc is the power voltage)  (8) 
     According to the invention of the present application, (2) the Q 1  turn-on loss, which accounts for about 40% of the total losses, can be eliminated. 
       FIG. 4  illustrates circuit efficiency in the switching power supply devices according to the related art and the invention. This graph shows the circuit efficiency for the output current Iout. Since it is possible to eliminate the turn-on loss of Q 1  under light load and set the dead time to the minimum under heavy load as stated above, the overall circuit efficiency can also be enhanced, by as much as approximately 8% when the output current Iout is 1 A. In other words, the output current Iout is small, efficiency can be enhanced by eliminating the aforementioned (2) turn-on loss of Q 1 . 
       FIG. 5  is a circuit diagram showing a switching power supply device, which is another preferred embodiment of the invention. In this embodiment, the input control circuit CONT and the logic circuit LOG of  FIG. 1  above are shown more specifically. Herein, the booster circuit comprising the boot strap capacitance CB and other elements is not shown. The PWM signal is supplied to one input each of an AND gate circuit G 2  and a NOR gate circuit G 3 . The output signal of the driver DV 2  driving the low potential side MOSFET Q 2  is inverted and inputted to the other input of the AND gate circuit G 2 . The output signal of the driver DV 1  driving the high potential side MOSFET Q 1  is inputted to the other input of the NOR gate circuit G 3 . The output signal HC of the AND gate circuit G 2  is conveyed to the input of the driver DV 1  through the level shift circuit LS and an AND gate circuit G 4 . Further, the output signal LC of the NOR gate circuit G 3  is conveyed to the input of the driver DV 2 . 
     This basically causes the MOSFET Q 1 , when the PWM signal is at a high level, to be turned on when the output signal of the driver DV 2  which turns off the MOSFET Q 2  is at a low level and the MOSFET Q 2 , when the PWM signal is at a low level, to be turned on when the output signal of the driver DV 1  which turns off the MOSFET Q 1  is at a low level. In this way, the basic dead time is set to a short period of monitoring the levels of the drivers D 1  and DV 2 . 
     In this embodiment, the detection signal of the voltage comparator circuit CMP is supplied to one input of a NOR gate circuit G 5 . The output signal of a delay circuit DLY which delays the output signal of an inverter circuit IV 1  which inverts the output signal of the driver DV 2  is supplied to the other input of the NOR gate circuit G 5 . The delay circuit DLY, constituting the permissible time setting circuit, limits the maximum dead time under light load. The output signal of the NOR gate circuit G 5  is supplied to one input of a NAND gate circuit G 6 . A mode signal MOD which is raised to a high level (logic 1) when under the light load is supplied to the other input of the NAND gate circuit G 6 . The output signal of this NAND gate circuit G 6  is used as a control signal for the AND gate circuit G 4  which conveys the drive signal for the high potential side MOSFET Q 1 . 
     When the detection signal of the voltage comparator circuit CMP and the input signal from the delay circuit DLY are at a low level (logic 0), the NOR gate circuit G 5  is outputting a high level (logic 1). Therefore, when the mode signal MOD is at a high level (logic 1), the NAND gate circuit G 6  creates a low level output signal. Therefore, even when the PWM signal is at a high level and moreover the output signal of the driver DV 2  which turns off the MOSFET Q 2  is at a low level as described above, the turning-on of the MOSFET Q 1  is stopped. When the detection signal of the voltage comparator circuit CMP varies to a high level, namely the midpoint voltage Vsw reaches about 80% or more of the input voltage Vin, the output signal of the NOR gate circuit G 5  varies to a low level. Accordingly, as the NAND gate circuit G 6  creates an output signal of a high level to open the gate of the AND gate circuit G 4 , the high potential side MOSFET Q 1  is turned on through the driver DV 1 . 
     If the detection signal the voltage comparator circuit CMP remains at a low level even after the lapse of the delay time of the delay circuit DLY, namely if the reverse current is too small to charge the parasitic capacitance sufficiently, the output signal of the delay circuit DLY will vary to a high level and causes the output signal of the NOR gate circuit G 5  to vary to a low level as described above. Therefore, as the NAND gate circuit G 6  creates an output signal of a high level to open the gate of the AND gate circuit G 4 , the high potential side MOSFET Q 1  is turned on through the driver DV 1 . 
     When the load of the load circuit (CPU or the like) is heavy, the mode signal MOD is brought down to a low level (logic 0). This causes the NAND gate circuit G 6  to output a high level irrespective of the output signals of the voltage comparator circuit CMP and the delay circuit DLY. Therefore, the high potential side MOSFET Q 1  is cause to create a control signal HC which achieves turning-on at the timing of the low level of the output signal of the driver DV 2  of the low potential side MOSFET Q 2 . This is intended to reduce losses in the body diode and thereby improve circuit efficiency under heavy load. 
       FIG. 6  is a schematic overall circuit diagram showing a switching power supply device, which is still another preferred embodiment of the invention. For this embodiment, the input voltage Vin is supposed to be a relatively high voltage, such as about 12 V and the output voltage Vout, a relatively low voltage, such as about 0.8 V, though the choice is not particularly limited to these. This output voltage Vout is used as the operational voltage for load circuits such as FPGA and CPU. 
     This embodiment is composed of the control circuit DVIC mounted with the input control circuit CONT, the level shift circuit LS and the drivers DV 1  and DV 2 , shown as a typical example in  FIG. 1  and  FIG. 5  above, together with a PWMIC for creating the PWM signal and unit components including the switch MOSFETs Q 1  and Q 2 , the inductor L 1 , the output capacitor Co and so forth. In order to control the output voltage Vout to a relatively low voltage, such as about 0.8 V, though not particularly limited to this, there is provided a voltage amplifier circuit comprising an operational amplifier OPA and resistances R 3  and R 4 . This voltage amplifier circuit constitutes a feedback control unit, which creates a voltage-amplified output voltage Vout′, such as the output voltage Vout×(R 3 +R 4 )/R 4 . This voltage Vout′ is divided by dividing resistances R 5  and R 6  in a ratio of R 6 /(R 5 +R 6 ), and the divided voltage is conveyed to a feedback terminal FB of the PWM control circuit PWMIC. 
     The feedback voltage conveyed to the feedback terminal FB is supplied to one input (−) of an error amplifier EA of the PWMIC. A band gap reference voltage Vref of about 1 V, though not particularly limited to this level, is supplied to the other input (+) of the error amplifier EA. The differential voltage between the feedback voltage and the reference voltage Vref is supplied to one input (−) of a voltage comparator circuit VC. A triangular wave created by a triangular wave generating circuit is supplied to the other input (+) of the voltage comparator circuit VC. The output signal of the voltage comparator circuit VC is inputted as a PWM signal to the input control circuit CONT provided in the driver DVIC. It is not particularly limited to a PWM signal, but what controls the output voltage Vout by regulating the switching of power MOSFETs, such as a pulse frequency modulation (PFM) signal or a pulse density modulation (PDM) signal, can be used as well. 
     Where a higher voltage than the band gap reference voltage Vref of about 1 V or so, such as 1.3 V, provided in the PWMIC is to be formed, as the output voltage Vout, the aforementioned voltage amplifier circuit comprising an operational amplifier OPA and resistances R 3  and R 4  can be dispensed with. By selectively disposing such voltage amplifier circuit comprising an operational amplifier OPA and resistances R 3  and R 4 , the output voltage Vout can be set in a broad range with the combination of circuits comprising DVIC, PWMIC and external components. 
     Although the invention accomplished by the present inventors have been hitherto described in specific terms with reference to preferred embodiments thereof, the invention is not limited to these embodiments, but can be modified in various manners without deviating from its essentials. For instance, the logic circuit LOG to validate/invalidate the detection signal of the voltage comparator circuit CMP can be realized in one or another of a number of specific configurations. Or, the mode signal MOD may be one using a signal such as a sleep mode or a standby mode of the load circuit CPU, or alternatively the switching power supply device itself may be provided with a circuit for detecting a light load state. This invention can be expensively used for voltage step-down type switching power supply devices.