Patent Publication Number: US-8120591-B2

Title: Detection of low noise frequencies for multiple frequency sensor panel stimulation

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 11/818,454, filed Jun. 13, 2007, the entire disclosure of which is incorporated herein by reference. 
    
    
     FIELD OF THE INVENTION 
     This relates to touch sensor panels used as input devices for computing systems, and more particularly, to the use of multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and to the use of multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel. 
     BACKGROUND OF THE INVENTION 
     Many types of input devices are presently available for performing operations in a computing system, such as buttons or keys, mice, trackballs, touch sensor panels, joysticks, touch screens and the like. Touch screens, in particular, are becoming increasingly popular because of their ease and versatility of operation as well as their declining price. Touch screens can include a touch sensor panel, which can be a clear panel with a touch-sensitive surface, and a display device that can be positioned behind the panel so that the touch-sensitive surface can substantially cover the viewable area of the display device. Touch screens can allow a user to perform various functions by touching the touch sensor panel using a finger, stylus or other object at a location dictated by a user interface (UI) being displayed by the display device. In general, touch screens can recognize a touch event and the position of the touch event on the touch sensor panel, and the computing system can then interpret the touch event in accordance with the display appearing at the time of the touch event, and thereafter can perform one or more actions based on the touch event. 
     Touch sensor panels can be formed from a matrix of row and column traces, with sensors or pixels present where the rows and columns cross over each other while being separated by a dielectric material. Each row can be driven by a stimulation signal, and touch locations can be identified because the charge injected into the columns due to the stimulation signal is proportional to the amount of touch. However, the high voltage that can be required for the stimulation signal can force the sensor panel circuitry to be larger in size, and separated into two or more discrete chips. In addition, touch screens formed from capacitance-based touch sensor panels and display devices such as liquid crystal displays (LCDs) can suffer from noise problems because the voltage switching required to operate an LCD can capacitively couple onto the columns of the touch sensor panel and cause inaccurate measurements of touch. Furthermore, alternating current (AC) adapters used to power or charge the system can also couple noise into the touchscreen. Other sources of noise can include switching power supplies in the system, backlight inverters, and light emitting diode (LED) pulse drivers. Each of these noise sources has a unique frequency and amplitude of interference that can change with respect to time. 
     SUMMARY OF THE INVENTION 
     This relates to the use of multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and to the use of multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel. Each of a plurality of sense channels can be coupled to a column in a touch sensor panel and can have multiple mixers. Each mixer in each sense channel can utilize a circuit capable of being controlled to generate a demodulation frequency of a particular frequency, phase and delay. 
     When performing a spectrum analyzer function, no stimulation signal is applied to any of the rows in the touch sensor panel. The sum of the output of all sense channels, which can represent the total charge being applied to the touch sensor panel including all detected noise, can be fed back to each of the mixers in each sense channel. The mixers can be paired up, and each pair of mixers can demodulate the sum of all sense channels using the in-phase (I) and quadrature (Q) signals of a particular frequency. The demodulated outputs of each mixer pair can be used to calculate the magnitude of the noise at that particular frequency, wherein the lower the magnitude, the lower the noise at that frequency. Several low noise frequencies can be selected for use in a subsequent touch sensor panel scan function. 
     When performing the touch sensor panel scan function, at each of multiple steps, various phases of the selected low noise frequencies can be used to simultaneously stimulate the rows of the touch sensor panel, and the multiple mixers in each sense channel can be configured to demodulate the signal received from the column connected to each sense channel using the selected low noise frequencies. The demodulated signals from the multiple mixers can then be saved. After all steps have been completed, the saved results can be used in calculations to determine an image of touch for the touch sensor panel at each frequency. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates an exemplary computing system that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to one embodiment of this invention. 
         FIG. 2   a  illustrates an exemplary mutual capacitance touch sensor panel according to one embodiment of this invention. 
         FIG. 2   b  is a side view of an exemplary pixel in a steady-state (no-touch) condition according to one embodiment of this invention. 
         FIG. 2   c  is a side view of an exemplary pixel in a dynamic (touch) condition according to one embodiment of this invention. 
         FIG. 3   a  illustrates a portion of an exemplary sense channel or event detection and demodulation circuit according to one embodiment of this invention. 
         FIG. 3   b  illustrates a simplified block diagram of N exemplary sense channel or event detection and demodulation circuits according to one embodiment of this invention. 
         FIG. 3   c  illustrates an exemplary block diagram of 10 sense channels that can be configured either as a spectrum analyzer or as panel scan logic according to one embodiment of this invention. 
         FIG. 4   a  illustrates an exemplary timing diagram showing an LCD phase and touch sensor panel phase according to one embodiment of this invention. 
         FIG. 4   b  illustrates an exemplary flow diagram describing the LCD phase and the touch sensor panel phase according to one embodiment of this invention. 
         FIG. 4   c  illustrates an exemplary capacitive scanning plan according to one embodiment of this invention. 
         FIG. 4   d  illustrates exemplary calculations for a particular channel M to compute full image results at different low noise frequencies according to one embodiment of this invention. 
         FIG. 5   a  illustrates an exemplary mobile telephone that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to one embodiment of this invention. 
         FIG. 5   b  illustrates an exemplary digital audio player that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to one embodiment of this invention. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     In the following description of preferred embodiments, reference is made to the accompanying drawings which form a part hereof, and in which it is shown by way of illustration specific embodiments in which the invention can be practiced. It is to be understood that other embodiments can be used and structural changes can be made without departing from the scope of the embodiments of this invention. 
     This relates to the use of multiple digital mixers to perform spectrum analysis of noise to identify low noise stimulation frequencies, and the use of multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel. Each of a plurality of sense channels can be coupled to a column in a touch sensor panel and can have multiple mixers. Each mixer in the sense channel can utilize a circuit capable of being controlled to generate a demodulation frequency of a particular frequency, phase and delay. 
     When performing a spectrum analyzer function, no stimulation signal is applied to any of the rows in the touch sensor panel. The sum of the output of all sense channels, which can represent the total charge being applied to the touch sensor panel including all detected noise, can be fed back to each of the mixers in each sense channel. The mixers can be paired up, and each pair of mixers can demodulate the sum of all sense channels using the in-phase (I) and quadrature (Q) signals of a particular frequency. The demodulated outputs of each mixer pair can be used to calculate the magnitude of the noise at that particular frequency, wherein the lower the magnitude, the lower the noise at that frequency. Several low noise frequencies can be selected for use in a subsequent touch sensor panel scan function. 
     When performing the touch sensor panel scan function, at each of multiple steps, various phases of the selected low noise frequencies can be used to simultaneously stimulate the rows of the touch sensor panel, and the multiple mixers in each sense channel can be configured to demodulate the signal received from the column connected to each sense channel using the selected low noise frequencies. The demodulated signals from the multiple mixers can then be saved. After all steps have been completed, the saved results can be used in calculations to determine an image of touch for the touch sensor panel at each frequency. 
     Although some embodiments of this invention may be described herein in terms of mutual capacitance touch sensors, it should be understood that embodiments of this invention are not so limited, but are generally applicable to other types of touch sensors such as self capacitance touch sensors. Furthermore, although the touch sensors in the touch sensor panel may be described herein in terms of an orthogonal array of touch sensors having rows and columns, it should be understood that embodiments of this invention are not limited to orthogonal arrays, but can be generally applicable to touch sensors arranged in any number of dimensions and orientations, including diagonal, concentric circle, and three-dimensional and random orientations. In addition, the touch sensor panel described herein can be either a single-touch or a multi-touch sensor panel, the latter of which is described in Applicant&#39;s co-pending U.S. application Ser. No. 10/842,862 entitled “Multipoint Touchscreen,” filed on May 6, 2004 and published as U.S. Published Application No. 2006/0097991 on May 11, 2006, the contents of which are incorporated by reference herein. 
       FIG. 1  illustrates exemplary computing system  100  that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to embodiments of the invention. Computing system  100  can include one or more panel processors  102  and peripherals  104 , and panel subsystem  106 . One or more panel processors  102  can include, for example, ARM968 processors or other processors with similar functionality and capabilities. However, in other embodiments, the panel processor functionality can be implemented instead by dedicated logic, such as a state machine. Peripherals  104  can include, but are not limited to, random access memory (RAM) or other types of memory or storage, watchdog timers and the like. Panel subsystem  106  can include, but is not limited to, one or more sense channels  108 , channel scan logic  110  and driver logic  114 . Channel scan logic  110  can access RAM  112 , autonomously read data from the sense channels and provide control for the sense channels. In addition, channel scan logic  110  can control driver logic  114  to generate stimulation signals  116  at various frequencies and phases that can be selectively applied to rows of touch sensor panel  124 . In some embodiments, panel subsystem  106 , panel processor  102  and peripherals  104  can be integrated into a single application specific integrated circuit (ASIC). 
     Touch sensor panel  124  can include a capacitive sensing medium having a plurality of row traces or driving lines and a plurality of column traces or sensing lines, although other sensing media can also be used. The row and column traces can be formed from a transparent conductive medium such as Indium Tin Oxide (ITO) or Antimony Tin Oxide (ATO), although other transparent and non-transparent materials such as copper can also be used. In some embodiments, the row and column traces can be perpendicular to each other, although in other embodiments other non-Cartesian orientations are possible. For example, in a polar coordinate system, the sensing lines can be concentric circles and the driving lines can be radially extending lines (or vice versa). It should be understood, therefore, that the terms “row” and “column,” “first dimension” and “second dimension,” or “first axis” and “second axis” as used herein are intended to encompass not only orthogonal grids, but the intersecting traces of other geometric configurations having first and second dimensions (e.g. the concentric and radial lines of a polar-coordinate arrangement). The rows and columns can be formed on a single side of a substantially transparent substrate separated by a substantially transparent dielectric material, on opposite sides of the substrate, or on two separate substrates separated by the dielectric material. 
     At the “intersections” of the traces, where the traces pass above and below (cross) each other (but do not make direct electrical contact with each other), the traces can essentially form two electrodes (although more than two traces could intersect as well). Each intersection of row and column traces can represent a capacitive sensing node and can be viewed as picture element (pixel)  126 , which can be particularly useful when touch sensor panel  124  is viewed as capturing an “image” of touch. (In other words, after panel subsystem  106  has determined whether a touch event has been detected at each touch sensor in the touch sensor panel, the pattern of touch sensors in the multi-touch panel at which a touch event occurred can be viewed as an “image” of touch (e.g. a pattern of fingers touching the panel).) The capacitance between row and column electrodes appears as a stray capacitance when the given row is held at direct current (DC) voltage levels and as a mutual signal capacitance Csig when the given row is stimulated with an alternating current (AC) signal. The presence of a finger or other object near or on the touch sensor panel can be detected by measuring changes to a signal charge Qsig present at the pixels being touched, which is a function of Csig. Each column of touch sensor panel  124  can drive sense channel  108  (also referred to herein as an event detection and demodulation circuit) in panel subsystem  106 . 
     Computing system  100  can also include host processor  128  for receiving outputs from panel processor  102  and performing actions based on the outputs that can include, but are not limited to, moving an object such as a cursor or pointer, scrolling or panning, adjusting control settings, opening a file or document, viewing a menu, making a selection, executing instructions, operating a peripheral device connected to the host device, answering a telephone call, placing a telephone call, terminating a telephone call, changing the volume or audio settings, storing information related to telephone communications such as addresses, frequently dialed numbers, received calls, missed calls, logging onto a computer or a computer network, permitting authorized individuals access to restricted areas of the computer or computer network, loading a user profile associated with a user&#39;s preferred arrangement of the computer desktop, permitting access to web content, launching a particular program, encrypting or decoding a message, and/or the like. Host processor  128  can also perform additional functions that may not be related to panel processing, and can be coupled to program storage  132  and display device  130  such as an LCD display for providing a UI to a user of the device. 
     In some systems, sensor panel  124  can be driven by high-voltage driver logic. The high voltages that can be required by the high-voltage driver logic (e.g. 18V) can force the high-voltage driver logic to be formed separate from panel subsystem  106 , which can operate at much lower digital logic voltage levels (e.g. 1.7 to 3.3V). However, in embodiments of the invention, on-chip driver logic  114  can replace the off-chip high voltage driver logic. Although panel subsystem  106  can have low, digital logic level supply voltages, on-chip driver logic  114  can generate a supply voltage greater that the digital logic level supply voltages by cascoding two transistors together to form charge pump  115 . Charge pump  115  can be used to generate stimulation signals  116  (Vstim) that can have amplitudes of about twice the digital logic level supply voltages (e.g. 3.4 to 6.6V). Although  FIG. 1  shows charge pump  115  separate from driver logic  114 , the charge pump can be part of the driver logic. 
       FIG. 2   a  illustrates exemplary mutual capacitance touch sensor panel  200  according to embodiments of the invention.  FIG. 2   a  indicates the presence of a stray capacitance Cstray at each pixel  202  located at the intersection of a row  204  and a column  206  trace (although Cstray for only one column is illustrated in  FIG. 2   a  for purposes of simplifying the figure). In the example of  FIG. 2   a , AC stimuli Vstim  214 , Vstim  215  and Vstim  217  can be applied to several rows, while other rows can be connected to DC. Vstim  214 , Vstim  215  and Vstim  217  can be at different frequencies and phases, as will be explained later. Each stimulation signal on a row can cause a charge Qsig=Csig×Vstim to be injected into the columns through the mutual capacitance present at the affected pixels. A change in the injected charge (Qsig_sense) can be detected when a finger, palm or other object is present at one or more of the affected pixels. Vstim signals  214 ,  215  and  217  can include one or more bursts of sine waves. Note that although  FIG. 2   a  illustrates rows  204  and columns  206  as being substantially perpendicular, they need not be so aligned, as described above. As described above, each column  206  can be connected to a sense channel (see sense channels  108  in  FIG. 1 ). 
       FIG. 2   b  is a side view of exemplary pixel  202  in a steady-state (no-touch) condition according to embodiments of the invention. In  FIG. 2   b , an electric field of electric field lines  208  of the mutual capacitance between column  206  and row  204  traces or electrodes separated by dielectric  210  is shown. 
       FIG. 2   c  is a side view of exemplary pixel  202  in a dynamic (touch) condition. In  FIG. 2   c , finger  212  has been placed near pixel  202 . Finger  212  is a low-impedance object at signal frequencies, and has an AC capacitance Cfinger from the column trace  204  to the body. The body has a self-capacitance to ground Cbody of about 200 pF, where Cbody is much larger than Cfinger. If finger  212  blocks some electric field lines  208  between the row and column electrodes (those fringing fields that exit the dielectric and pass through the air above the row electrode), those electric field lines are shunted to ground through the capacitance path inherent in the finger and the body, and as a result, the steady state signal capacitance Csig is reduced by ΔCsig. In other words, the combined body and finger capacitance act to reduce Csig by an amount ΔCsig (which can also be referred to herein as Csig_sense), and can act as a shunt or dynamic return path to ground, blocking some of the electric fields as resulting in a reduced net signal capacitance. The signal capacitance at the pixel becomes Csig−ΔCsig, where Csig represents the static (no touch) component and ΔCsig represents the dynamic (touch) component. Note that Csig−ΔCsig may always be nonzero due to the inability of a finger, palm or other object to block all electric fields, especially those electric fields that remain entirely within the dielectric material. In addition, it should be understood that as a finger is pushed harder or more completely onto the multi-touch panel, the finger can tend to flatten, blocking more and more of the electric fields, and thus ΔCsig can be variable and representative of how completely the finger is pushing down on the panel (i.e. a range from “no-touch” to “full-touch”). 
       FIG. 3   a  illustrates a portion of exemplary sense channel or event detection and demodulation circuit  300  according to embodiments of the invention. One or more sense channels  300  can be present in the panel subsystem. Each column from a touch sensor panel can be connected to sense channel  300 . Each sense channel  300  can include virtual-ground amplifier  302 , amplifier output circuit  309  (to be explained in greater detail below), signal mixer  304 , and accumulator  308 . Note that amplifier output circuit  309  can also be connected to other signal mixers and associated circuitry not shown in  FIG. 3   a  to simplify the figure. 
     Virtual-ground amplifier  302 , which can also be referred to as a DC amplifier or a charge amplifier, can include feedback capacitor Cfb and feedback resistor Rfb. In some embodiments, because of the much smaller amount of charge that can be injected into a row due to lower Vstim amplitudes, Cfb can be made much smaller than in some previous designs. However, in other embodiments, because as many as all rows can be simultaneously stimulated at the same time, which tends to add charge, Cfb is not reduced in size. 
       FIG. 3   a  shows, in dashed lines, the total steady-state signal capacitance Csig_tot that can be contributed by a touch sensor panel column connected to sense channel  300  when one or more input stimuli Vstim are applied to one or more rows in the touch sensor panel and no finger, palm or other object is present. In a steady-state, no-touch condition, the total signal charge Qsig_tot injected into the column is the sum of all charge injected into the column by each stimulated row. In other words, Qsig_tot=Σ(Csig*Vstim for all stimulated rows). Each sense channel coupled to a column can detect any change in the total signal charge due to the presence of a finger, palm or other body part or object at one or more pixels in that column. In other words, Qsig_tot_sense=Σ((Csig−Csig_sense)*Vstim for all stimulated rows). 
     As noted above, there can be an inherent stray capacitance Cstray at each pixel on the touch sensor panel. In virtual ground charge amplifier  302 , with the + (noninverting) input tied to reference voltage Vref, the − (inverting) input can also be driven to Vref, and a DC operating point can be established. Therefore, regardless of how much Csig is present at the input to virtual ground charge amplifier  302 , the − input can always be driven to Vref. Because of the characteristics of virtual ground charge amplifier  302 , any charge Qstray that is stored in Cstray is constant, because the voltage across Cstray is kept constant by the charge amplifier. Therefore, no matter how much stray capacitance Cstray is added to the − input, the net charge into Cstray will always be zero. The input charge is accordingly zero when the corresponding row is kept at DC and is purely a function of Csig and Vstim when the corresponding row is stimulated. In either case, because there is no charge across Csig, the stray capacitance is rejected, and it essentially drops out of any equations. Thus, even with a hand over the touch sensor panel, although Cstray can increase, the output will be unaffected by the change in Cstray. 
     The gain of virtual ground amplifier  302  can be small (e.g. 0.1) and can be computed as the ratio of Csig_tot and feedback capacitor Cfb. The adjustable feedback capacitor Cfb can convert the charge Qsig to the voltage Vout. The output Vout of virtual ground amplifier  302  is a voltage that can be computed as the ratio of −Csig/Cfb multiplied by Vstim referenced to Vref. The Vstim signaling can therefore appear at the output of virtual ground amplifier  302  as signals having a much smaller amplitude. However, when a finger is present, the amplitude of the output can be even further reduced, because the signal capacitance is reduced by ΔCsig. The output of charge amplifier  302  is the superposition of all row stimulus signals multiplied by each of the Csig values on the column associated with that charge amplifier. A column can have some pixels which are driven by a frequency at positive phase, and simultaneously have other pixels which are driven by that same frequency at negative phase (or 180 degrees out of phase). In this case, the total component of the charge amplifier output signal at that frequency can be the amplitude and phase associated with the sum of the product of each of the Csig values multiplied by each of the stimulus waveforms. For example, if two rows are driven at positive phase, and two rows are driven at negative phase, and the Csig values are all equal, then the total output signal will be zero. If the finger gets near one of the pixels being driven at positive phase, and the associated Csig reduces, then the total output at that frequency will have negative phase. 
     Vstim, as applied to a row in the touch sensor panel, can be generated as a burst of sine waves (e.g. sine waves with smoothly changing amplitudes in order to be spectrally narrow) or other non-DC signaling in an otherwise DC signal, although in some embodiments the sine waves representing Vstim can be preceded and followed by other non-DC signaling. If Vstim is applied to a row and a signal capacitance is present at a column connected to sense channel  300 , the output of charge amplifier  302  associated with that particular stimulus can be sine wave train  310  centered at Vref with a peak-to-peak (p-p) amplitude in the steady-state condition that can be a fraction of the p-p amplitude of Vstim, the fraction corresponding to the gain of charge amplifier  302 . For example, if Vstim includes 6.6V p-p sine waves and the gain of the charge amplifier is 0.1, then the output of the charge amplifier associated with this row can be approximately 0.67V p-p sine wave. In should be noted that the signal from all rows are superimposed at the output of the preamp. The analog output from the preamp is converted to digital in block  309 . The output from  309  can be mixed in digital signal mixer  304  (which is a digital multiplier) with demodulation waveform Fstim  316 . 
     Because Vstim can create undesirable harmonics, especially if formed from square waves, demodulation waveform Fstim  316  can be a Gaussian sine wave that can be digitally generated from numerically controlled oscillator (NCO)  315  and synchronized to Vstim. It should be understood that in addition to NCOs  315 , which are used for digital demodulation, independent NCOs can be connected to digital-to-analog converters (DACs), whose outputs can be optionally inverted and used as the row stimulus. NCO  315  can include a numerical control input to set the output frequency, a control input to set the delay, and a control input to enable the NCO to generate an in-phase (I) or quadrature (Q) signal. Signal mixer  304  can demodulate the output of charge amplifier  310  by subtracting Fstim  316  from the output to provide better noise rejection. Signal mixer  304  can reject all frequencies outside the passband, which can in one example be about +/−30 kHz around Fstim. This noise rejection can be beneficial in noisy environment with many sources of noise, such as 802.11, Bluetooth and the like, all having some characteristic frequency that can interfere with the sensitive (femtofarad level) sense channel  300 . For each frequency of interest being demodulated, signal mixer  304  is essentially a synchronous rectifier as the frequency of the signal at its inputs is the same, and as a result, signal mixer output  314  is essentially a rectified Gaussian sine wave. 
       FIG. 3   b  illustrates a simplified block diagram of N exemplary sense channel or event detection and demodulation circuits  300  according to embodiments of the invention. As noted above, each charge amplifier or programmable gain amplifier (PGA)  302  in sense channel  300  can be connected to amplifier output circuit  309 , which in turn can be connected to R signal mixers  304  through multiplexer  303 . Amplifier output circuit  309  can include anti-aliasing filter  301 , ADC  303 , and result register  305 . Each signal mixer  304  can be demodulated with a signal from a separate NCO  315 . The demodulated output of each signal mixer  304  can be connected to a separate accumulator  308  and results register  307 . 
     It should be understood that PGA  302 , which may have detected a higher amount of charge generated from a high-voltage Vstim signal (e.g. 18V) in previous designs, can now detect a lower amount of charge generated from a lower voltage Vstim signal (e.g. 6.6V). Furthermore, NCOs  315  can cause the output of charge amplifier  302  to be demodulated simultaneously yet differently, because each NCO  310  can generate signals at different frequencies, delays and phases. Each signal mixer  304  in a particular sense channel  300  can therefore generate an output representative of roughly one-Rth the charge of previous designs, but because there are R mixers, each demodulating at a different frequency, each sense channel can still detect about the same total amount of charge as in previous designs. 
     In  FIG. 3   b , signal mixers  304  and accumulators  308  can be implemented digitally instead of in analog circuitry inside an ASIC. Having the mixers and accumulators implemented digitally instead of in analog circuitry inside the ASIC can save about 15% in die space. 
       FIG. 3   c  illustrates an exemplary block diagram of 10 sense channels  300  that can be configured either as a spectrum analyzer or as panel scan logic according to embodiments of the invention. In the example of  FIG. 3   c , each of 10 sense channels  300  can be connected to a separate column in a touch sensor panel. Note that each sense channel  300  can include multiplexer or switch  303 , to be explained in further detail below. The solid-line connections in  FIG. 3   c  can represent the sense channels configured as panel scan logic, and the dashed-line connections can represent the sense channels configured as a spectrum analyzer.  FIG. 3   c  will be discussed in greater detail hereinafter. 
       FIG. 4   a  illustrates exemplary timing diagram  400  showing LCD phase  402  and the vertical blanking or touch sensor panel phase  404  according to embodiments of the invention. During LCD phase  402 , the LCD can be actively switching and can be generating voltages needed to generate images. No panel scanning is performed at this time. During touch sensor panel phase  404 , the sense channels can be configured as a spectrum analyzer to identify low noise frequencies, and can also be configured as panel scan logic to detect and locate an image of touch. 
       FIG. 4   b  illustrates exemplary flow diagram  406  describing LCD phase  402  and touch sensor panel phase  404  corresponding to the example of  FIG. 3   c  (the present example) according to embodiments of the invention. In Step  0 , the LCD can be updated as described above. 
     Steps  1 - 3  can represent a low noise frequency identification phase  406 . In Step  1 , the sense channels can be configured as a spectrum analyzer. The purpose of the spectrum analyzer is to identify several low noise frequencies for subsequent use in a panel scan. With no stimulation frequencies applied to any of the rows of the touch sensor panel, the sum of the output of all sense channels, which represent the total charge being applied to the touch sensor panel including all detected noise, can be fed back to each of the mixers in each sense channel. The mixers can be paired up, and each pair of mixers can demodulate the sum of all sense channels using the in-phase (I) and quadrature (Q) signals of a particular frequency. The demodulated outputs of each mixer pair can be used to calculate the magnitude of the noise at that particular frequency, wherein the lower the magnitude, the lower the noise at that frequency. 
     In Step  2 , the process of Step  1  can be repeated for a different set of frequencies. 
     In Step  3 , several low noise frequencies can be selected for use in a subsequent touch sensor panel scan by identifying those frequencies producing the lowest calculated magnitude value. 
     Steps  4 - 19  can represent a panel scan phase  408 . In Steps  4 - 19 , the sense channels can be configured as panel scan logic. At each of Steps  4 - 19 , various phases of the selected low noise frequencies can be used to simultaneously stimulate the rows of the touch sensor panel, and the multiple mixers in each sense channel can be configured to demodulate the signal received from the column connected to each sense channel using the selected low noise frequencies. The demodulated signals from the multiple mixers can then be saved. 
     In Step  20 , after all steps have been completed, the saved results can be used in calculations to determine an image of touch for the touch sensor panel at each of the selected low noise frequencies. 
     Referring again to the present example as shown in  FIG. 3   c , while sense channels  300  are configured as a spectrum analyzer, no stimulation signal is applied to any of the rows in the touch sensor panel. In the present example, there are 10 columns and therefore 10 sense channels  300 , and three mixers  304  for each sense channel  300 , for a total of 30 mixers. The outputs of all amplifier output circuits  309  in every sense channel  300  can be summed together using summing circuit  340 , and fed into all mixers  304  through multiplexer or switch  303 , which can be configured to select the output of summing circuit  340  instead of charge amplifier  302 . 
     While the sense channels are configured as a spectrum analyzer, the background coupling onto the columns can be measured. Because no Vstim is applied to any row, there is no Csig at any pixel, and any touches on the panel should not affect the noise result (unless the touching finger or other object couples noise onto ground). By adding all outputs of all amplifier output circuits  309  together in adder  340 , one digital bitstream can be obtained representing the total noise being received into the touch sensor panel. The frequencies of the noise and the pixels at which the noise is being generated are not known prior to spectrum analysis, but do become known after spectrum analysis has been completed. The pixel at which the noise is being generated is not known and is not recovered after spectrum analysis, but because the bitstream is being used as a general noise collector, they need not be known. 
     While configured as a spectrum analyzer, the 30 mixers in the example of  FIG. 3   c  can be used in 15 pairs, each pair demodulating the I and Q signals for 15 different frequencies as generated by NCOs  315 . These frequencies can be between 200 kHz and 300 kHz, for example. NCOs  315  can produce a digital rampsine wave that can be used by digital mixers  304  to demodulate the noise output of summing circuit  340 . For example, NCO  315 _ 0 _A can generate the I component of frequency F 0 , while NCO  315 _ 0 _B can generate the Q component of F 0 . Similarly, NCO  315 _ 0 _C can generate the I component of frequency F 1 , NCO  315 _ 1 _A can generate the Q component of F 1 , NCO  315 _ 1 _B can generate the I component of frequency F 2 , NCO  315 _ 1 _C can generate the Q component of F 2 , etc. 
     The output of summing circuit  340  (the noise signal) can then be demodulated by the I and Q components of F 0  through F 14  using the 15 pairs of mixers. The result of each mixer  304  can be accumulated in accumulators  308 . Each accumulator  308  can be a digital register that, over a sample time period, can accumulate (add together) the instantaneous values from mixer  304 . At the end of the sample time period, the accumulated value represents the amount of noise signal at that frequency and phase. 
     The accumulated results of an I and Q demodulation at a particular frequency can represent the amount of content at that frequency that is either in phase or in quadrature. These two values can then be used in magnitude and phase calculation circuit  342  to find the absolute value of the total magnitude (amplitude) at that frequency. A higher magnitude can mean a higher background noise level at that frequency. The magnitude value computed by each magnitude and phase calculation circuit  342  can be saved. Note that without the Q component, noise that was out of phase with the demodulation frequency can remain be undetected. 
     This entire process can be repeated for 15 different frequencies F 15 -F 29 . The saved magnitude values for each of the 30 frequencies can then be compared, and the three frequencies with the lowest magnitude values (and therefore the lowest noise levels), referred to herein as frequencies A, B and C, can be chosen. In general, the number of low noise frequencies chosen can correspond to the number of mixers in each sense channel. 
     Still referring to  FIG. 3   c , when sense channels  300  are configured as panel scan logic, the dashed lines in  FIG. 3   c  can be ignored. At each of Steps  4 - 19 , various phases of the selected low noise frequencies can be used to simultaneously stimulate the rows of the touch sensor panel, and the multiple mixers in each sense channel can be configured to demodulate the signal received from the column connected to each sense channel using the selected low noise frequencies A, B and C. In the example of  FIG. 3   c , NCO_ 0 _A can generate frequency A, NCO_ 0 _B can generate frequency B, NCO_ 0 _C can generate frequency C, NCO_ 1 _A can generate frequency A, NCO_ 1 _B can generate frequency B, NCO_ 1 _C can generate frequency C, etc. The demodulated signals from each mixer  304  in each sense channel can then be accumulated in accumulators  308 , and saved. 
     In general, the R mixer outputs for any sense channel M (where M=0 to N−1) demodulated by R low noise frequencies F 0 , F 1  . . . F R-1  can be represented by the notation xF 0 S[chM], xF 1 S[chM] . . . xF R-1 S[chM], where xF 0  represents the output of a mixer demodulated with frequency F 0 , xF 1  represents the output of a mixer demodulated with frequency F 1 , xF R-1  represents the output of a mixer demodulated with frequency F R-1 , and S represents the sequence number in the panel scan phase. 
     Therefore, in Step  4  (representing sequence number  1  in the panel scan phase), and using low noise frequencies A, B and C as the demodulation frequencies, the outputs to be saved can be referred to as xa 1 [ch 0 ], xb 1 [ch 0 ], xc 1 [ch 0 ], xa 1 [ch  1 ], xb 1 [ch 1 ], xc 1 [ch 1 ], . . . xa 1 [ch 9 ], xb 1 [ch 9 ], xc 1 [ch 9 ]. Thus, in the present example, 30 results are saved in Step  4 . In Step  5  (representing sequence number  2  in the panel scan phase), the 30 results to be saved can be referred to as xa 2 [ch 0 ], xb 2 [ch 0 ], xc 2 [ch 0 ], xa 2 [ch 1 ], xb 2 [ch 1 ], xc 2 [ch 1 ], . . . xa 2 [ch 9 ], xb 2 [ch 9 ], xc 2 [ch 9 ]. The 30 outputs to be saved in each of Steps  6 - 19  can be similarly named. 
     It should be understood that the additional logic outside the sense channels in  FIG. 3   c  can be implemented in the channel scan logic  110  of  FIG. 1 , although it could also be located elsewhere. 
       FIG. 4   c  illustrates an exemplary capacitive scanning plan  410  corresponding to the present example according to embodiments of the invention.  FIG. 4   c  describes Steps  0 - 19  as shown in  FIG. 4   b  for an exemplary sensor panel having 15 rows R 0 -R 14 . 
     Step  0  can represent the LCD phase at which time the LCD can be updated. The LCD phase can take about 12 ms, during which time no row can be stimulated. 
     Steps  1 - 19  can represent the vertical blanking interval for the LCD, during which time the LCD is not changing voltages. 
     Steps  1 - 3  can represent the low noise frequency identification phase which can take about 0.6 ms, again during which time no row can be stimulated. In Step  1 , the I and Q components of different frequencies ranging from 200 kHz to 300 kHz (separated by at least 10 kHz) can be simultaneously applied to pairs of mixers in the sense channels configured as a spectrum analyzer, and a magnitude of the noise at those frequencies can be saved. In Step  2 , the I and Q components of different frequencies ranging from 300 kHz to 400 kHz can be simultaneously applied to pairs of mixers in the sense channels configured as a spectrum analyzer, and a magnitude of the noise at those frequencies can be saved. In Step  3 , the lowest noise frequencies A, B and C can be identified by locating the frequencies that produced the lowest saved magnitudes. The identification of the lowest noise frequencies can be done solely on the measured spectra measured in steps  1  and  2 , or it can also take into account historical measurements from steps  1  and  2  of previous frames. 
     Steps  4 - 19  can represent the panel scan phase which can take about 3.4 ms. 
     In Step  4 , which can take about 0.2 ms, positive and negative phases of A, B and C can be applied to some rows, while other rows can be left unstimulated. It should be understood that +A can represent scan frequency A with a positive phase, −A can represent scan frequency A with a negative phase, +B can represent scan frequency B with a positive phase, −B can represent scan frequency B with a negative phase, +C can represent scan frequency C with a positive phase, and −C can represent scan frequency C with a negative phase. The charge amplifiers in the sense channels coupled to the columns of the sensor panel can detect the total charge coupled onto the column due to the rows being stimulated. The output of each charge amplifier can be demodulated by the three mixers in the sense channel, each mixer receiving either demodulation frequency A, B or C. Results or values xa 1 , xb 1  and xc 1  can be obtained and saved, where xa 1 , xb 1  and xc 1  are vectors. For example, xa 1  can be a vector with 10 values xa 1 [ch 0 ], xa 1 [ch 1 ], xa 1 [ch 2 ] . . . xa 1 [ch 9 ], xb 1  can be a vector with 10 values xb 1 [ch 0 ], xb 1 [ch 1 ], xb 1 [ch 2 ] . . . xb 1 [ch 9 ], and xc 1  can be a vector with 10 values xc 1 [ch 0 ], xc 1 [ch 1 ], xc 1  [ch 2 ] . . . xc 1  [ch 9 ]. 
     In particular, in Step  4 , +A is applied to rows  0 ,  4 ,  8  and  12 , +B, −B, +B and −B are applied to rows  1 ,  5 ,  9  and  13 , respectively, +C, −C, +C and −C are applied to rows  2 ,  6 ,  10  and  14 , respectively, and no stimulation is applied to rows  3 ,  7 ,  11  and  15 . The sense channel connected to column  0  senses the charge being injected into column  0  from all stimulated rows, at the noted frequencies and phases. The three mixers in the sense channel can now be set to demodulate A, B and C, and three different vector results xa 1 , xb 1  and xc 1  can be obtained for the sense channel. Vector xa 1 , for example, can represent the sum of the charge injected into columns  0 - 9  at the four rows being stimulated by +A (e.g. rows  0 ,  4 ,  8  and  12 ). Vector xa 1  does not provide complete information, however, as the particular row at which a touch occurred is still unknown. In parallel, in the same Step  4 , rows  1  and  5  can be stimulated with +B, and rows  9  and  13  can be stimulated with −B, and vector xb 1  can represent the sum of the charge injected into columns  0 - 9  at the rows being stimulated by +B and −B (e.g. rows  1 ,  5 ,  9  and  13 ). In parallel, in the same Step  4 , rows  2  and  14  can be stimulated with +C, and rows  6  and  10  can be stimulated with −C, and vector xc 1  can represent the sum of the charge injected into columns  0 - 9  at the rows being stimulated by +C and −C (e.g. rows  2 ,  6 ,  10  and  14 ). Thus, at the conclusion of Step  4 , three vectors containing 10 results each, for a total of 30 results, are obtained and stored. 
     Steps  5 - 19  are similar to Step  4 , except that different phases of A, B and C can be applied to different rows, and different vector results are obtained at each step. At the conclusion of Step  19 , a total of 480 results will have been obtained in the example of  FIG. 4   c . By obtaining the 480 results at each of Steps  4 - 19 , a combinatorial, factorial approach is used wherein incrementally, for each pixel, information is obtained regarding the image of touch for each of the three frequencies A, B and C. 
     It should be noted that Steps  4 - 19  illustrate a combination of two features, multi-phase scanning and multi-frequency scanning. Each feature can have its own benefit. Multi-frequency scanning can save time by a factor of three, while multi-phase scanning can provide a better signal-to-noise ratio (SNR) by about a factor of two. 
     Multi-phase scanning can be employed by simultaneously stimulating most or all of the rows using different phases of multiple frequencies. Multi-phase scanning is described in Applicant&#39;s co-pending U.S. application Ser. No. 11/619,433 entitled “Simultaneous Sensing Arrangement,” filed on Jan. 3, 2007, the contents of which are incorporated by reference herein. One benefit of multi-phase scanning is that more information can be obtained from a single panel scan. Multi-phase scanning can achieve a more accurate result because it minimizes the possibility of inaccuracies that can be produced due to certain alignments of the phases of the stimulation frequency and noise. 
     In addition, multi-frequency scanning can be employed by simultaneously stimulating most or all of the rows using multiple frequencies. As noted above, multi-frequency scanning saves time. For example, in some previous methods, 15 rows can be scanned in 15 steps at frequency A, then the 15 rows can be scanned in 15 steps at frequency B, then the 15 rows can be scanned in 15 steps at frequency C, for a total of 45 steps. However, using multi-frequency scanning as shown in the example of  FIG. 4   c , only a total of 16 steps (Steps  4  through Step  19 ) can be required. Multi-frequency in its simplest embodiment can include simultaneously scanning R 0  at frequency A, R 1  at frequency B, and R 2  at frequency C in a first step, then simultaneously scanning R 1  at frequency A, R 2  at frequency B, and R 3  at frequency C in step  2 , etc. for a total of 15 steps. 
     At the conclusion of Steps  4 - 19 , when the 480 results described above have been obtained and stored, additional calculations can be performed utilizing these 480 results. 
       FIG. 4   d  illustrates exemplary calculations for a particular channel M to compute full image results at different low noise frequencies corresponding to the present example according to embodiments of the invention. In the present example, for each channel M, where M=0 to 9, the 45 computations shown in  FIG. 4   d  can be performed to obtain a row result for each row and each frequency A, B and C. Each set of 45 computations for each channel can generate a resultant pixel value for the column of pixels associated with that channel. For example, the Row  0 , frequency A computation (xa 1 [chM]+xa 2 [chM]+xa 3 [chM]+xa 4 [chM])/4 can generate the row  0 , channel M result for frequency A. In the present example, after all computations have been performed and stored for every channel, a total of 450 results will have been obtained. These computations correspond to Step  20  of  FIG. 4   b.    
     Of these 450 results, there will be 150 for frequency A, 150 for frequency B, and 150 for frequency C. The 150 results for a particular frequency represent an image map or image of touch at that frequency because a unique value is provided for each column (i.e. channel) and row intersection. These touch images can then be processed by software that synthesizes the three images and looks at their characteristics to determine which frequencies are inherently noisy and which frequencies are inherently clean. Further processing can then be performed. For example, if all three frequencies A, B and C are all relatively noise-free, the results can be averaged together. 
     It should be understood that the computations shown in  FIGS. 4   c  and  4   d  can be performed under control of panel processor  102  or host processor  128  of  FIG. 1 , although they could also be performed elsewhere. 
       FIG. 5   a  illustrates an exemplary mobile telephone  536  that can include touch sensor panel  524 , display device  530  bonded to the sensor panel using pressure sensitive adhesive (PSA)  534 , and other computing system blocks in computing system  100  of  FIG. 1  that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to embodiments of the invention. 
       FIG. 5   b  illustrates an exemplary digital audio/video player  540  that can include touch sensor panel  524 , display device  530  bonded to the sensor panel using pressure sensitive adhesive (PSA)  534 , and other computing system blocks in computing system  100  of  FIG. 1  that can utilize multiple digital mixers to perform spectrum analysis of noise and identify low noise stimulation frequencies, and can utilize multiple stimulation frequencies and phases to detect and localize touch events on a touch sensor panel according to embodiments of the invention. 
     Although embodiments of this invention have been fully described with reference to the accompanying drawings, it is to be noted that various changes and modifications will become apparent to those skilled in the art. Such changes and modifications are to be understood as being included within the scope of embodiments of this invention as defined by the appended claims.