Patent Publication Number: US-8115574-B2

Title: Low pass filter with embedded resonator

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates generally to low pass filters for microwave signals. More particularly, it relates to providing improved frequency characteristics in the microwave spectrum for such filters. 
     2. Description of Related Art 
     The microwave portion of the spectrum, usually defined as extending from roughly 300 MHz to about 300 GHz, is used for wireless signals among various devices such as, for example, cellular telephones, personal digital assistants (PDAs), WiFi devices, and navigational systems. 
     Because many different devices concurrently use the microwave spectrum, government regulations and various agreements have divided it into discrete spectrum bands, which are often further split into smaller sub-bands, thereby minimizing interference. To meet such regulations and agreements, and to meet communication quality requirements, transmitting devices are generally prohibited from emitting energy over a specified level outside of their assigned bands and, preferably, receiving devices are constructed to limit receipt of energy to only their assigned bands 
     Various microwave filters are therefore incorporated into transmitters and receivers, to limit their broadcast and receipt of signals, respectively, to particular frequencies. For this reason, the performance qualities of the microwave filters often have significant effect on the quality of communications and, further, are a determining factor for spacing between channels and, hence, the usable capacity of the spectrum. 
     Microwave filters may be configured to have low pass (LPF), band pass (BPF) or high pass (HPF) characteristics, each typically having at least one pass band, transition band and stop band. 
     For purposes of brevity this disclosure, however, will describe various exemplary embodiments and arrangements in reference to microwave LPFs. This is simply to focus the description on the novel features and aspects of the invention, to better enable persons of ordinary skill in the art to make and use it based on this disclosure. However, otherwise stated or clear from the context, the invention and all of its various embodiments may be readily practiced in alternative arrangements as microwave BPFs and/or HPFs simply by, for example, applying conventional filter design methods to translate or reconfigure the disclosed microwave LPFs to microwave BPFs or HPFs. 
     As known to persons skilled in the relevant arts, an ideal microwave LPF blocks all frequencies above a given cut-off frequency, has a zero-width transition band, and passes without attenuation all signal frequencies below the cut-off. 
     Realizable microwave LPFs, however, do not have such characteristics. Realizable microwave LPFs have pass band attenuation, meaning that some of desired signal energy is lost, a finite attenuation, meaning that some undesired signal energy gets through, and a slope-like transition band extending from the cut-off frequency to the reject band. Therefore, among the various measures of microwave LPF transmission quality, three are: stop-band attenuation, band-pass loss, and cut-off slope. 
     One well-known type of microwave LPF is the stepped-impedance resonator (SIR) filter, which comprises a succession of resonant sections, each section having a high impedance subsection that steps to a low impedance subsection. The resonant sections may be configured in various ways, such as coaxial, microstrip, or strip line. 
       FIG. 1  is a three-dimensional view of an exemplar coaxial SIR LPF  10  according to the related art, with its outer conductor removed for clarity. 
     As shown in  FIG. 1 , a traditional coaxial SIR LPF  10  may comprise a series of N resonator sections, each referenced as  12   n , n=1 to N. Each section  12   n  comprises a low impedance subsection  14   n  followed by a high impedance subsection  16   n  which, at microwave frequencies, embody a capacitor and an inductor, respectively. Each section  12   n  therefore forms an inductor-capacitor (LC) resonator. 
       FIG. 2  shows a lumped parameter model  20  for a coaxial SIR LPF such as the  FIG. 1  exemplar  10 . 
     Referring to  FIG. 2 , lumped parameter model  20  depicts an SIR LPF such as the  FIG. 1  example  10 , as comprising N resonator sections  22   n , each having an inductor element L n  and a capacitor element C n , each having a respective reactance value corresponding, in reference to  FIG. 1 , to the impedance of its modeled subsection  14   n  and  16   n . The relative values of L n  and C n , each set by physical parameters such as width, length and materials, in turn set the resonant frequency of each set  22   n . Therefore, an appropriate LPF characteristic may be obtained by selecting appropriate dimensions and materials for each section  12   n . 
       FIG. 3  shows an illustrative frequency response  30 , based on an example seven-pole related art SIR LPF such as, for example, the  FIG. 1  exemplar  10 . Referring to  FIG. 3 , the example frequency response  30  has an example upper “cut-off” frequency, labeled  32 , at approximately 5 GHz. The 5 GHz value in this example is arbitrary, but the form of the frequency response is representative of a related art seven-pole SIR LPF. The slope of the frequency response  34  above the example 5 GHz cut-off, labeled  32 , is not very sharp. This is shown particularly by the attenuation  36  of only approximately 12 dB at approximately 5.5 GHz. Spurious modes may appear at 5.5 GHz, through, due to harmonics, or integral multiples of the resonant sections (not shown in  FIG. 3 ) that form the SIP LPF. 
     There are known methods directed to solving the problem of spurious bands. All of these methods, however, have shortcomings. 
     For example, one method is to add another LPF, such as a mask filter, to the SIR LPF. This has drawbacks, though, including increased cost and, particularly, pass-band insertion loss. Further, adding a mask filter in line with a main filter may increase the complexity of the tuning procedure of the overall microwave system. 
     Another method is the addition of an arrangement of inductors, as described by U.S. Pat. No. 2,641,646 to Thomas. However, the method taught by Thomas may have many of the some of the same shortcomings as using an additional LPF. In addition, Thomas may require the use of heavy wire or copper tubing, materials that may not be appropriate for a low cost microwave LPF microwave cavity. 
     Another related method directed to solving the problem of spurious modes is taught by Published U.S. Patent Application No. 2003/0001697 to Bennett et al. Bennet teaches intermediate suppression elements, interspersed within the SIR structure. However, this method may require complete reconfiguration of the SIR filter structure 
     SUMMARY OF THE INVENTION 
     Accordingly, a need exists for a simply structured, easy to manufacture SIR LPF that has built-in suppression of close to pass band spurious signals. This invention and its various described exemplary embodiments are directed to this need and provide, with various other features and benefits, a SIR LPF having an embedded notch frequency resonator filter with a simple, easy to manufacture structure, readily implementing substantially any practical specification requirement for a spurious-free LPF. 
     According to one aspect of one or more embodiments, the embedded notch resonator filter may be formed by an inner conductor, integrated with a multi-pole filter such as an SIR-LPF, having a simple, integral structure that supports a dielectric spacer. This support structure and the dielectric spacer may, in arrangement with a face of a distal end of a transmission line, form a capacitive couple, of capacitance CC, coupling to a capacitance CR in parallel with an inductance LR, terminating to an effective ground, forming an LC resonator. 
     One aspect of one or more of the various exemplary embodiments includes a coaxial SIR LPF that has an inner conductor extending from a succession of resonant cavity sections, the inner conductor having at one distal end a projecting structure that supports a dielectric spacer having a gap thickness GP, the dielectric spacer abutting a distal end of a center conductor of a transmission line, to form the capacitance CR and inductance LR of an LC resonator, wherein CR is based, at least in part, on the gap thickness GP. 
     One aspect of one or more of the various exemplary embodiments includes an SIR LPF having a first center conductor that has, near one distal end, a step-down shoulder and a projection that extends a distance LN from the step-down shoulder to the distal end, a dielectric spacer with a hollow cylindrical portion surrounding the projection, and a flange, having a thickness GP, abutting the step-down shoulder, and a second center conductor with a distal end having a bore, arranged such that the hollow cylindrical portion of the dielectric spacer surrounding the projection extends into the bore, to form the capacitance CR and inductance LR of an LC resonator, wherein LR is based, at least in part, on the length LN. 
     According to another aspect of the various exemplary embodiments, the bore extends to a well-bottom surface in the second center conductor, the dielectric spacer includes an end wall at a distal end of the hollow cylindrical portion of the dielectric spacer, such that an annular face surrounding the bore at a distal end of the second inner conductor is spaced the gap distance GP by the flange from the step-down shoulder of the first center conductor, and the terminal end of the projection is spaced, by the end wall of the dielectric spacer, from the well-bottom surface of the recess. 
     According to another aspect of the various exemplary embodiments, simply varying the length LN of the projection varies the center frequency of the resonant notch frequency filter. 
     According to another aspect of the various exemplary embodiments, simply varying the length gap GP varies the maximum attenuation without significant change of the center frequency of the resonant notch frequency filter. 
     According to another aspect of the various exemplary embodiments, the second center conductor may be a distal end of a conventional coaxial transmission line, having a conventional center conductor readily drilled, machined, or otherwise formed by, for example, conventional tools, to have a recess with a diameter and length to accommodate the projection and the cylindrical portion of the dielectric spacer. 
     According to another aspect of the various exemplary embodiments, multiple sections of the resonant notch frequency filter may be cascaded together, to provide a wider stop band of desired rejection, and thereby attenuate multiple spurious modes. 
     The above-summarized objects, aspect and advantages of the invention and its various exemplary are only illustrative of those that can be achieved by the various exemplary embodiments, and are not intended to be exhaustive or limiting. These and other objects, aspects and advantages of the various exemplary embodiments will be apparent from the description herein, or can be learned from practicing the various exemplary embodiments, both as embodied herein or as modified in view of any variation which may be apparent to those skilled in the art. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       To better understand various exemplary embodiments, reference is made to the accompanying drawings, wherein: 
         FIG. 1  is a three-dimensional view of a coaxial SIR LPF according to the related art, with the outer conductor removed for clarity; 
         FIG. 2  shows a lumped parameter model for a SIR LPF according to the related art; 
         FIG. 3  shows a frequency response diagram for a seven-pole LPF according to the related art; 
         FIG. 4A  is a three-dimensional depiction of one example SIR LPF with embedded resonator notch filter according to one embodiment; 
         FIG. 4B  is an enlargement of a cross-section of the example embedded resonator notch filter portion of the  FIG. 4A  example; 
         FIG. 5  is a further enlargement of the  FIG. 4B  example, showing one example gap and projection length; 
         FIG. 6A  shows a lumped element model of an example embedded resonator portion according to various embodiments; 
         FIG. 6B  shows a distributed model of an example embedded resonator portion according to various embodiments of the present invention; 
         FIG. 7  is an illustration of one example frequency response obtainable from an embedded resonator implementing a one-pole resonator, according to various exemplary embodiments; 
         FIG. 8  is an illustration of one example frequency response obtainable from an example according to one embodiment, comprising an SIR LPF having an example embedded one pole resonator such as the  FIG. 7  example; 
         FIG. 9  shows one aspect according to various exemplary embodiments, of varying the notch frequency of the embedded one pole resonator portion by varying the gap which, according to one example, varies a CR value of the aspect&#39;s achieved LC resonator; 
         FIG. 10  shows one aspect according to various exemplary embodiments, of varying the notch frequency of the embedded one pole resonator by varying a length parameter which, according to one example, varies an LR value of the aspect&#39;s achieved LC resonator; 
         FIG. 11  is an illustration of one example frequency response obtainable from an embedded resonator implementing a two-pole resonator, according to various exemplary embodiments; and 
         FIG. 12  is an illustration of one example frequency response obtainable from one example, according to one embodiment, comprising an SIR LPF having an example embedded two pole resonator such as the  FIG. 11  example. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS OF THE INVENTION 
     Referring now to the drawings, in which like numerals refer to like components or steps, there are disclosed broad aspects of various exemplary embodiments. 
     In one broad aspect, a subject of this invention is an embedded resonator that may be integrated with various filter structures such as, for example, a coaxial SIR LPF. According to aspects having an SIR LPF, the embedded notch resonator introduces finite transmission zeros to the all transmission-pole response of the coaxial low-pass filter, which significantly enhances the spurious suppression of the coaxial filter. This provides an integrated filter/notch resonator having, among other features, sharp rejection near the operating band of the system, while maintaining a wide spurious suppression window. 
       FIG. 4A  is a three-dimensional depiction of one example  40  having an SIR LPF  42  with an embedded resonator having structure including a capacitive coupling at region  44 , according to one embodiment.  FIG. 4B  is an enlargement of a cross-section of portion  44 . 
     Referring to  FIG. 4B , a first transmission  46  is formed with a projection  46 A, which extends into a bore (not separately labeled in  FIG. 4B ) formed in a second, abutting transmission line  48 . Referring to  FIG. 4A , in the depicted example the second transmission line is the distal end of an inner conductor extending from the SIR LPF  42 . A dielectric spacer  50 , having a flange portion  50 A and a cylindrical sleeve portion  50 B separates the projection  46 A from the bore in  48 , and separates the shoulder (not separately labeled in  FIG. 4B ) where the projection  46 A extends from the transmission line  46  by a gap G from the opposite annular ring face (not separately labeled in  FIG. 4B ) of the second transmission line  48 . In the  FIG. 4B  example, the dielectric spacer has an end wall  50 C that separates a terminal end (not separately numbered in  FIG. 4B ) of the projection  46 A from a well-bottom of the bore in the second transmission line  46 . The thickness (not separately labeled in  FIG. 4B ) of the end wall  50 C and the thickness of the walls (not separately labeled in  FIG. 4B ) of the cylindrical sleeve portion  50 B are preferably, but are not necessarily, approximately the same thickness as G. 
     As will be understood, and as explained in greater detail, opposing surfaces of the projection  46 A and the bore in transmission line  48 , and of the shoulder on line  46  with the annular face of transmission line  48 , form an LC resonator.  FIGS. 6A and 6B , described in greater detail in later sections, show a lumped-model and a distributed model, respectively, of the LC resonator. are 
       FIG. 5  shows a further cross-sectional view of an example embedded resonator  500 , generally structured according to  FIG. 4B . The  FIG. 5  example  500 , however, in comparison to  FIG. 4B  has a reverse orientation as to which transmission line has the projection and which has the accommodating bore and, therefore, is separately numbered. 
     Referring to  FIG. 5 , the depicted example  500  comprises a first transmission line having  50 A having, at its distal end, a projection  50 B extending a length LN from a shoulder  50 C. The projection has a diameter D 1 . The second transmission line  52  has, at its distal end facing the distal end  50 A of the first transmission line, a bore surface  52 A extending approximately LN from a annular face  52 B at the extreme distal end of the line  52  to a well-bottom face  52 C. The diameter of the bore  52 B (not separately labeled) is preferably such that the cylindrical gap G 1  existing between the outer surface of the projection  50 B and the bore surface  52 A is approximately the same as the gap GP separating the shoulder face  50 C of the first transmission line from the annular face  52 B of the second transmission line  52 . Further, the extending length LN of the projection  50 B is preferably such that the gap G 2  separating the distal end of the projection  50 B from the well-bottom face  52 C of the bore is approximately the same as the gap GP. 
     With continuing reference to  FIG. 5 , a dielectric spacer (not collectively labeled) has a flange portion  54 A of approximately thickness GP filling the space between the shoulder face  50 C of the first transmission line  50  and the annular face  52 B of the second transmission line. The dielectric spacer includes a sleeve portion  54 B, having a thickness approximately equal to G 1 , surrounding the hollow cylindrical space between the outer surface of the projection  50 B and the bore surface  52 A. and has an end wall  54 C within the space G 2  separating the distal end of the projection  50 B from the well-bottom face  52 C of the bore. 
       FIG. 6A  is a lumped element model  60 A of an example embedded resonator according to one disclosed embodiment such as, for example, a structure as exemplified at  FIG. 5 . 
     Referring to  FIGS. 5 and 6A , capacitance CR and inductance LR model as a parallel LC resonator the reactive impedance along the path of the shoulder  50 C, the projection  50 B, separated from the bore  52 B by the dielectric spacer, and the capacitance CC models the coupling capacitance between the junction of the first transmission line  50 B and the second transmission line  52  and the LC resonator. The length LN substantially sets the inductance LR, and the GP, G 1  and G 2  substantially set the capacitance CR. Therefore, as readily seen by persons skill in the art, the notch frequency is easily set. 
       FIG. 6B  is a distributed model  60 B of an example embedded resonator according to one disclosed embodiment such as, for example, a structure as exemplified at  FIG. 5 . 
       FIG. 7  is an illustration of one example frequency response obtainable from one example according to one embodiment, comprising one example SIR LPF having an example embedded one pole resonator, such as that achieved by the  FIG. 5  structure, having LN and GP, G 1  and G 2  values selected for suppressing one spurious at  72  which, in the depicted example, is 5 GHz. 
     As seen at the plot section  74  of the S11 parameter shown in  FIG. 7 , the frequency response of the one pole resonator according to the  FIG. 5  example embodiments has a sharp drop just above 5 GHz, increasing to a magnitude of almost −40 dB at the 5 GHz center frequency. Plot line  76  represents the S21 transmission parameter. The  FIG. 7  frequency response is readily obtainable on a structure according to that depicted at  FIG. 5 , by selecting GP and LN dimensions based on this disclosure, using conventional computer modeling and design methods well known to persons of ordinary skill in the art. 
       FIG. 8  is an illustration of one example frequency response obtainable from an example according to one embodiment, comprising an SIR LPF such as modeled at  FIG. 3 , having an example embedded one pole resonator according to the invention, such as the  FIG. 7  example. 
     As seen from the plot sections  82  and  84  of the S11 parameter shown in  FIG. 8 , compared to the  FIG. 3  frequency response for the same SIR LPF, this embodiment provides substantially improved cut-off slope, including rejection of spurious mode signals occurring just above the operating frequency, e.g., 5 GHz at, with only a single pole implementation, a spurious signal suppression that exceeds −50 dB. Plot line  86  represents the S21 transmission parameter. 
       FIG. 9  shows one aspect according to various exemplary embodiments, of varying the maximum attenuation at the notch frequency of an embedded one pole, such as achieved by a structure as illustrated at  FIG. 5 , by varying the gap GP, G 1  and G 2  labeled in  FIG. 5 . This varies the coupling capacitance CC, and the resonant LC capacitance CR shown, for example, in the lumped parameter model at  FIG. 6A . In the example variation of the gap GP shown in  FIG. 9 , an example LN value was fixed at about 1.2,″ setting a resonant frequency of roughly 4.5 GHz. Varying values of GP, using example 0.01″, 0.015″, and 0.025,″ labeled  92 ,  94  and  96 , respectively, provides significant variation of the attenuation. 
     Example ranges and values of GP depend on various factors, including frequency requirements, environment, cost and manufacturability. For example, a square coaxial line may have an outer width of, for example, 0.235″ and an inner diameter of, for example, 0.109″. In such a case, the smallest practical gap, meaning easily manufactured with controllable quality, would have a dimension of about 0.01″. Referring to  FIGS. 5 and 6A , the LR value could then be adjusted, by setting LN, to fine tune the frequency response. 
       FIG. 10  shows one aspect according to various exemplary embodiments, of varying the notch frequency of an embedded one pole resonator, such as achieved by a structure as illustrated at  FIG. 5 , by varying the projection length labeled in  FIG. 5  as LN which, as described above, varies the LR value of the aspect&#39;s achieved LC shown, for example, in the lumped parameter model at  FIG. 6A . 
     Referring to  FIG. 10 , the lowest resonant frequency  102 , obtained by setting LN=1.2″, centered at only 4.6 GHz. Progressively higher resonant frequencies, labeled  104 ,  106  and  108 , were obtained by decreasing LN to 1.0″, 0.8″, and 0.6″, resulting in frequencies of 5.5 GHz, 6.7 GHz, and 8.6 GHz, respectively. This illustrates that the present embodiments provide not only a simple structure, but ready adjustment of resonant frequency by varying just one simple structural parameter, namely, the length LN of the projection. Thus, one may increase the central frequency of resonator  500  of  FIG. 5  significantly by gradually decreasing the L value of resonator  500 . 
       FIG. 11  is an illustration of one example frequency response obtainable from an embedded resonator implementing a two-pole resonator, according to various exemplary embodiments. Two poles are achieved by cascading two structures according to the embodiments, such as shown at FIG.  5 , with appropriate gap GP and length LN values. Because this resonator embodiment has two poles, as opposed to the single pole of the resonator exhibiting the  FIG. 8  response, the spurious stop band, such as labeled  112 , is a wider band. As also shown, The magnitude of spurious mode suppression may, for example, have a magnitude of about −50 dB. 
       FIG. 12  is an illustration of one example frequency response obtainable from one example, according to one embodiment, comprising an SIR LPF, such as the sample represented at  FIG. 3 , having an example embedded two pole resonator such as the  FIG. 11  example. 
     As seen the two pole resonator provides a large rejection band  122  above 5 GHz. The magnitude of spurious mode suppression may be as great as −60 dB for this embodiment. 
     Although the various exemplary embodiments have been described in detail with particular reference to certain exemplary aspects thereof, it should be understood that the invention is capable of other different embodiments, and its details are capable of modifications in various obvious respects. 
     For example, a plurality of embedded resonators such as shown at  FIG. 5  could be integrated with an existing coaxial SIP LPF to realize a LPF function with finite transmission zeros. 
     Further, as can be readily seen by persons skilled in the relevant art, conventional microwave transformers may be inserted between the embedded resonators according to the disclosed embodiments, to provide a desired return loss characteristic. 
     As is readily apparent to those skilled in the art, variations and modifications can be affected while remaining within the spirit and scope of the invention. Accordingly, the foregoing disclosure, description, and figures are for illustrative purposes only, and do not in any way limit the invention, which is defined only by the claims.