Patent Publication Number: US-6987428-B2

Title: Electromagnetic coupler flexible circuit with a curved coupling portion

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   This application is a continuation of U.S. patent application Ser. No. 09/714,899, filed on Nov. 15, 2000, now U.S. Pat. No. 6,573,801, issued Jun. 3, 2003. 

   FIELD OF INVENTION 
   This invention is related to the field of electromagnetic coupling devices for bus communication. 
   BACKGROUND OF THE INVENTION 
   Electromagnetic coupling devices enable energy to be transferred between components of a system via interacting electric and magnetic fields. These interactions are quantified using coupling coefficients. The capacitive coupling coefficient is the ratio of the per unit length coupling capacitance, C m , to the geometric mean of the per unit length capacitances of the two coupled lines, C l . Similarly, the inductive coupling coefficient is the ratio of the per unit length mutual inductance, L m , to the geometric mean of the per unit length inductances of the two coupled lines, L l . 
     FIG. 1  shows a conventional broadside coupler in a space having x, y, and z orientations, where the two broadest faces of two adjacent printed circuit board conductor lines A and B are electromagnetically coupled having a distance D in between.  FIG. 2  shows an edge coupler in a space having x, y, and z orientations, where the narrow faces of two conductors A and B on the same layer are coupled having a distance D in between. 
   Conventional coupling devices suffer from deficiencies in several areas. The coupling devices exhibit significant variations in the capacitive coupling coefficient due to manufacturing tolerances in the line geometry and in the relative position of the two coupled lines (“x,y,z variations”). Furthermore, in common manufacturing practices, the width of conductors is subject to variations of between +/−0.5 and +/−1.0 mils, the relative alignment of conductor layers within a printed circuit board (PCB) is subject to variations of +/−5 mils (x,y axis), the distance between conductor layers can vary by +/−2 mils (z axis), and the location of holes for guide pins is subject to +/−4 mil variations (x,y axis). Therefore, conventional couplers are too sensitive to misalignment to be used in computer systems. 
   The present invention addresses these and other deficiencies of conventional couplers. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like references indicate similar elements, and in which: 
       FIG. 1  shows a prior art broadside coupler. 
       FIG. 2  shows a prior art edge coupler. 
       FIGS. 3 ,  4 , and  5  show embodiments of a portion of a coupler including two conductors. 
       FIGS. 6A ,  6 B, and  7  show embodiments of multiple crossed coupler segments. 
       FIGS. 8 and 9  show variations in capacitive coupling coefficient. 
       FIGS. 10A and 10B  show embodiments of a coupler. 
       FIGS. 11A and 11B  show a digital bus communication system having multiple couplers. 
       FIGS. 12A ,  12 B,  12 C, and  12 D show embodiments of a cross-section of a coupler. 
       FIG. 13  shows another embodiment of a cross-section of a coupler. 
       FIG. 14  shows an orthogonal view of the cross-section shown in FIG.  13 . 
       FIG. 15  shows an embodiment of a coupler on a motherboard and a flex circuit. 
   

   DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION 
   An electromagnetic (EM) coupler is disclosed. For one embodiment, the EM coupler includes a first transmission structure having a first geometry and a second transmission structure having a second geometry, which may be different than the first geometry. An EM coupling is formed between the first and second transmission structures. For one embodiment, the first and second geometries are selected to reduce sensitivity of EM coupling to relative positions of the first and second transmission structures. The EM coupler structure may be physically separated into two component halves to be used in an interconnect application. 
   For one embodiment, the EM coupler provides a broadband coupling device that is separable, bi-directional, and provides robust performance despite misalignment of the transmission structures. The coupler may further have an impedance that is controlled over a wide frequency range to prevent losses from reflections. Thus, the coupler may be used to transmit and receive digital signals. 
   For one embodiment, the EM coupler also provides bidirectional signal transfer; i.e. the transmission properties of the coupler are essentially the same in the forward and reverse signal transfer directions. For one embodiment, the line impedance of the EM coupler is compatible with the circuitry of a computer system. 
     FIG. 3  shows a coupler in a space having x, y, and z orientations that includes an arrangement of sections of two conductors A and B separated by a dielectric such as air, for example, represented by a distance D.  FIG. 4  shows a top view of the sections of the conductors having x and y orientations. As shown in  FIG. 4 , conductor A is rotated by an angle  410  from the common longitudinal axis, while conductor B is rotated by an equal but opposite angle  410  from the same common longitudinal axis. 
     FIG. 5  shows a coupler having a total capacitance that includes a parallel plate capacitance and a fringe capacitance. In overlapping area  510 , the capacitance contribution from the overlapping sections of the conductors is generally similar to that of a parallel plate capacitor with parallelogram shaped plates. The capacitance between the conductors A and B in regions  520  is a fringe capacitance. The outer bounding edges  525  show the points where the added fringe capacitance between the two conductors A and B becomes negligible, e.g. less than 0.1%, of the total capacitance of the coupler. 
   The combination of the parallel plate capacitance and the fringe capacitance provides a nearly constant coupling capacitance in the face of deviations from a nominal position. This constant coupling capacitance provides robust coupling even if the conductors are misaligned. Therefore, the two conductors can be moved relative to each other in the x and y directions, without a significant change in their mutual capacitive coupling coefficient. 
   This constant coupling coefficient behavior under x, y translation holds providing that the lengths of the two conductors are such that no disturbing feature, such as, for example, the end of either conductor or a bend in either of the conductors, falls into the overlapping area  510  or the fringe regions  520  of the conductors in such a manner as to significantly perturb the parallel plate and the fringe capacitance contributions. However, if a disturbing feature is present, the coupler may still function, but the coupling coefficient may change significantly and the performance may be degraded. 
   Referring to  FIGS. 3 ,  4 , and  5 , if the vertical separation distance d (see  FIG. 3 ) between the two conductors is increased, the contribution of the parallel plate component in the region  510  in  FIG. 5  decreases as a function of 1/d. However, the fringe capacitance in regions  520  of  FIG. 5  can contribute as much as 25% of the total coupling capacitance between the conductors. The distance between surface elements of the conductors in the fringe capacitance regions is determined by both the conductor separation distance d (see  FIG. 3 ) and the selected angle  410  (see FIG.  4 ). The fringe capacitance contribution changes at a rate significantly less than 1/d. The rate of change in the coupling coefficient between conductors A and B, as shown in  FIG. 5 , separated by a distance d (see  FIG. 3 ) and rotated by a selected angle  410  (see FIG.  4 ), is therefore significantly less than the rate of change between couplers having broadside or edge configurations, as shown in  FIGS. 1 and 2 , where nearly all of the coupling capacitance shows a 1/d dependency. 
   The coupling coefficient may be increased by the use of multiple crossed coupler segments for a fixed length of coupler region as shown in FIG.  6 A. Referring to  FIG. 6A , a conductor A has been formed from multiple connected segments lying in a plane, where adjacent segments are arranged with an alternating angular displacement about the longitudinal axis of the conductor. A second similarly segmented conductor B is separated from conductor A by a dielectric at some predetermined distance, with its segments lying in a plane parallel to that of conductor A and arranged so that the angular displacement of its segments are in the opposite sense to the corresponding segments in conductor A, to form the zig-zag structure shown in FIG.  6 A. The structures from conductor A and conductor B have their longitudinal axes aligned collinearly in their nominal position, as shown in FIG.  6 A. (Alternatively, one conductor may have a zig-zag geometry, and the other conductor may have a straight line geometry. This alternative embodiment is shown in  FIG. 6B , which shows a coupler having one straight conductor A, and also another conductor B, which is segmented in a zig-zag geometry, including parallel plate capacitance regions  610  and fringe capacitance regions  620 .) 
   By providing a number of parallel plate capacitance regions  610  and fringe capacitance regions  620  per unit length, the geometry shown in  FIG. 6A  increases the capacitive coupling coefficient available between the coupled conductors A and B, while retaining the alignment insensitivity characteristics of the coupler shown in FIG.  5 . 
   In addition to the capacitive coupling coefficient, the coupler also has an inductive coupling coefficient, which is derived from the mutual inductance between the conductors and the self inductance of each conductor. The mutual inductance describes the energy that is magnetically transferred from one conductor to the other. For example, a time-varying electric current flowing through one conductor generates a time-varying magnetic field which causes an electric current to flow through the other conductor. The self inductance describes the energy that is stored when an electric current flows through a conductor and generates a magnetic field. 
   The inductive coupling coefficient, which is the ratio of the mutual inductance between the conductors to the geometric mean of the self inductance of each individual conductor, is also proportional to the geometric mean distance between the conductors. The mutual inductance is proportional to the length of the coupler conductors. The capacitive and inductive parameters of a structure with a given geometry are determined by the material properties of the structure. Therefore, once a structure has been designed with an appropriate geometry to obtain a desired set of capacitive parameters, the inductive parameters are also determined. 
   The interaction of the capacitive and inductive coupling characteristics becomes significant, especially at higher frequencies. This interaction results in directivity for the coupler. By controlling the length of the coupler to be a preferred fraction of a wavelength at a desired lower frequency, the relative magnitude of energy flow in the forward and reverse directions on the receiving conductor of the coupler (directivity) is determined over a preferred frequency range. For example 1 cm of length provides approximately 3 dB directivity over a frequency range of 400 megahertz (MHz) to 3 gigahertz (GHz). 
   The magnitude of the coupling coefficient for the coupler shown in  FIG. 6A  remains substantially unchanged over a large range of relative x and y displacements of the conductors A and B as long as the distance between the adjacent edges of the two conductors is greater than a given distance. In the limiting case shown in  FIG. 7 , an increase in the coupling coefficient begins to occur when the x, y displacement becomes sufficiently large to bring the adjacent edges  710  and  720  of the conductors A and B into close proximity. The range of x,y displacements for which the coupling coefficient remains essentially constant is therefore controlled by selection of an appropriate segment length, such as 0.125 cm for example, and an appropriate displacement angle, such as 35 degrees, for example. Further, by selection of appropriate values for the conductor widths, conductor separation and number of segments, a range of coupling coefficients may be obtained. 
   For example,  FIG. 8  shows the computed variation in capacitive coupling coefficient for a coupler composed of 5 mil wide conductors. The x and y dimension offsets in  FIG. 8  are up to 8 mils. In this range, the variation in the capacitive coupling coefficient is less than +/−2% about the average. 
     FIG. 9  shows the computed variation in capacitive coupling coefficient with a change in the separation distance between the coupler conductors in the z axis. It shows that for a +/−30% change in conductor separation, the capacitive coupling coefficient varies by less than +/−15%. This compares with parallel plate based geometries shown in  FIGS. 1 and 2  which show a +40/−30% variation over the same range of conductor separations. 
   In addition to the stability of the coupling coefficients of the geometry shown in  FIG. 6A , several alternative geometries may be used in the coupler structure. These alternative geometries may reduce far-field electromagnetic radiation, increase broadband behavior of the coupler, reduce impedance discontinuities, and enable the use of alternate materials for improved performance and flexibility. 
   One embodiment of an alternative geometry for the EM coupler is shown in FIG.  10 A. Referring to  FIG. 10A , the EM coupler includes a differential pair of conductors  1010  and  1012 . Conductor  1010  is coupled to a second conductor  1014 , while conductor  1012  is coupled to a second conductor  1016 . A first reference plane  1019  is placed below the first set of conductors  1010 ,  1012 , to act as a return conductor for these transmission lines. A second reference plane  1020  is placed above the second set of conductors  1014  and  1016  to act as a return conductor for the transmission lines  1014  and  1016 . Ends  1010 B and  1012 B of the first conductors  1010  and  1012  are terminated with matched termination resistors  1024  and  1026 . Ends  1014 B and  1016 B of the second set of conductors are also terminated with matched resistors  1028  and  1030 . 
   A differential digital signal is applied to ends  1010 A and  1012 A of the first conductors, and a resulting differential coupled signal is then observed at the set of conductor ends  1014 A and  1016 A. Conversely, a differential digital signal is applied to ends  1014 A and  1016 A of the second conductors, and a resulting differential coupled signal is then observed at the set of conductor ends  1010 A and  1012 A. Thus, the first and second set of conductors are reciprocally coupled by their electromagnetic fields. Alignment insensitivity of the coupler aids differential signaling by reducing mismatches between the coupler formed by conductors  1010  and  1014  and the coupler formed by conductors  1012  and  1016 . 
   The differential coupler shown in  FIG. 10A  reduces the effects of radiation. The use of differential signaling, with anti-phased currents flowing in the differential conductor pair, causes the radiation to fall rapidly to zero as the distance from the differential pair is increased. The differential signaling version of the coupler therefore offers lower far-field electromagnetic radiation levels than a single ended implementation. In addition to this differential embodiment, the coupler may be used in a single ended implementation, where a single conductor couples electromagnetically to a single conductor, as shown in FIG.  6 A. 
   In addition, the effects of far-field radiation may be further reduced by selecting an even number of conductor segments (e.g., eight segments) for the coupler. Thus offers potentially lower far field electromagnetic radiation levels compared to an implementation using an odd number of conductor segments. 
   The structure of  FIG. 10A , which couples the differential signals, has a differential pair of conductors that alternately approach each other and then turn away. Because the conductors  1014  and  1016  of the second transmission structure have segments with equal and opposite angular displacements to conductors  1010  and  1012 , respectively, this structure reduces the effects of capacitive crosstalk between conductors  1010  and  1016  and conductors  1012  and  1014  due to misalignment from X,Y variation of the conductors. 
     FIG. 10B  shows an alternative geometry to the embodiment of Figure  10 A. In  FIG. 10B , the pair of differential conductors  1010  and  1012  have a segmented, angular rotated structure. Each segment of one conductor from the pair has an angular displacement such that the segment is parallel to a corresponding segment of the other conductor of the pair of conductors. This results in a differential pair where the conductors maintain parallel positions to each other throughout the length of the coupler. In this configuration, the conductors  1014  and  1016  of the second transmission structure have segments with equal and opposite angular displacements to conductors  1010  and  1012 , respectively, while also keeping corresponding segments of conductors  1014  and  1016  parallel to each other. However, this alternative embodiment of  FIG. 10B  is subject to greater sensitivity to capacitive crosstalk than the embodiment of FIG.  10 A. 
   For one embodiment, the coupler is designed to avoid impedance discontinuities, or changes in the electromagnetic field structure, by not using connections between multiple printed circuit board (PCB) layers, and avoiding abrupt (right angle) bends. (However, in an alternative embodiment, a coupler may be designed with discontinuities or changes in field structure.) The discontinuity effects of the small angular bends in between the coupler segments is further reduced by chamfering the outer edge of the bend slightly to keep the conductor width reasonably constant throughout the bend. 
     FIG. 11A  represents electrical properties of an embodiment of a system that includes multiple couplers in a digital bus communications system. A conductor  1112 , which may be on the motherboard of a computer, for example, incorporates two or more couplers  1140 ,  1141  along its length. The end  1112 A of the conductor  1112  on the motherboard is connected to a transceiver  1110  to permit the transmission or reception of digital signals in a bi-directional manner. The end  1112 B of the conductor  1112  on the motherboard is terminated with a resistor  1136  equal to the impedance of the conductor. 
   The ends  1114 B and  1134 B of each coupled conductor are terminated with matching resistors  1130 ,  1132  for high frequency operation, the ends  1114 B and  1134 B are selected to be the ends furthest from the motherboard transceiver  1110 , because of signal directionality. Each daughter card has a transceiver  1120 ,  1122  connected to the end of the coupled conductor  1114 A,  1134 A, respectively. The transceiver  1110  transmits digital data which is received via the couplers  1140 ,  1141  by the daughter card transceivers  1120 ,  1122 . Conversely, transceivers  1120 ,  1122  may separately transmit data through couplers  1140 ,  1141  for reception and decoding at transceiver  1110 .  FIG. 11B  shows a differential version of the multiple couplers for a bus communication system, where the transceivers  1120  and  1122  are coupled to the conductor  1112  via a respective coupler. 
   This embodiment includes a data channel, such as a bus  1112 , having substantially uniform electrical properties for transferring signals among devices that are coupled through the data channel. The uniform electrical properties are supported by an electromagnetic coupling scheme that allows higher frequency signaling to be employed without significantly increasing noise attributable to transmission line effects. This is achieved by ensuring that only a small amount of energy (e.g., less than 1%) is transferred between the bus and the coupled daughter card. A preferred embodiment of this system is constructed in such a way that daughter cards containing devices  1120  and  1122  may be removed from or inserted to the system with little effect on the communication bandwidth of the bus. 
     FIG. 12A  shows an embodiment of a cross-section of the coupler of  FIG. 10A , shown at the point where the conductors cross. A differential pair of conductive signal traces  1230 A and  1230 B are coupled with another differential pair of conductive signal traces,  1236 A and  1236 B. Dielectric  1212  separates conductive signal traces  1230 A and  1230 B. Dielectric  1220  separates conductive signal traces  1236 A and  1236 B. Dielectric  1216  separates the differential pairs. Conductive reference planes  1210  and  1222  provide return paths for the conductive signal traces. The coupler may be constructed as an integral part of the computer motherboard. The conductive components  1230 A,  1230 B,  1236 A,  1236 B of the coupler with selected width (e.g., 5 mils) and thickness (e.g., 1.4 mils) may be constructed using conventional etching techniques on the surface of a dielectric sheet  1216 . The sheet  1216  may have a preferred thickness (e.g., 3.5 mils) and dielectric constant (e.g., 4.5). Additional dielectric layers  1212  and  1220 , with preferred thickness (e.g., 12 mils) and dielectric constant are added to provide the required spacing between the coupler elements  1230 A,  1230 B,  1236 A,  1236 B and the outer conductive reference planes  1210 ,  1222 . The end connections to the motherboard coupled conductors can then be connected to the daughter card using conventional impedance controlled electrical connectors as is currently common practice. 
   By placing cross-coupled conductors of the coupler between upper and lower conductive reference planes,  1210  and  1222 , as shown in  FIG. 12A , a dual stripline structure is formed. Stripline structures have the same even mode propagation velocity (the velocity for the wave propagation mode between the conductors and the reference planes) as the odd mode propagation velocity (the velocity of the wave propagation mode between the individual conductors of the coupler). This results in broadband behavior, allowing the coupler to operate up to frequencies in the microwave region. 
   Alternatively, the coupler may include a microstrip reference plane, a coplanar reference plane, or may have no reference plane at all. One alternative embodiment is shown in  FIG. 12B , which shows the two pairs of conductors  1230 A,  1230 B and  1236 A,  1236 B separated in a dielectric medium with no reference planes. This structure will form an EM coupler, however, it is not particularly suited for impedance control or wide bandwidth characteristics. 
     FIG. 12C  shows a microstrip configuration for the coupler with both pairs of conductors  1230 A,  1230 B, and  1236 A,  1236 B referenced to a single reference plane  1222 . This microstrip embodiment improves the impedance and bandwidth characteristics over that of FIG.  12 B. Alternatively, a coplanar waveguide structure of  FIG. 12D  may be constructed with reference conductors  1210  and  1222  in the same plane as the corresponding conductive signal lines  1230 A,  1230 B and  1236 A,  1236 B. 
   The dielectrics in  FIGS. 12A through 12D  may be any dielectric material, for example air or FR 4 . The bandwidth may be improved by selecting dielectric materials with similar dielectric constants. In  FIGS. 12A through 12D , conductors  1230 A and  1230 B may have a different width than conductors  1236 A and  1236 B. Also, dielectric  1212  may have a different thickness than dielectric  1220 . 
   A separable embodiment of the coupler of  FIG. 10A  is exemplified in the cross-sectional view of FIG.  13 . In this embodiment, motherboard conductors  1336 A and  1336 B are constructed on the outer layers  1360  of a printed circuit card, with a width such as 8 mils for example, and a thickness of 2.1 mils for example. The daughter-board conductors  1330 A and  1330 B are contained in a flexible circuit  1350 , which is pressed onto the surface of the motherboard. The conductors  1330 A and  1330 B may be 10 mils wide and 0.7 mils thick, for example. In  FIG. 13 , conductive reference plane  1322  is an internal power or ground plane as commonly used in printed circuit motherboards. The dielectric layer  1320  with preferred thickness and dielectric constant (e.g., 5 mils and 4.5, respectively) is used to provide the correct spacing between the motherboard conductive signal traces  1336 A,  1336 B and the conductive reference plane  1322 . 
   The outer surface of the board may be coated with a thin dielectric coating or solder mask  1318 , although this is not essential to the operation of the coupler. The daughter card portion of the coupler is provided with a conductive reference plane  1310  attached to the top surface of a flexible dielectric  1312  with preferred thickness (e.g., 2 mils) and dielectric constant (e.g., 4.5). The daughter card conductive signal traces  1330 A,  1330 B are constructed on the lower surface of the flexible dielectric  1312 . A dielectric adhesive  1314  is used to attach a dielectric or cover-lay film  1316  with preferred thickness (e.g., 0.5 mils) and dielectric constant (e.g., 3.8). The required coupling coefficient is achieved by selecting the preferred thicknesses and dielectric constants for the dielectric  1316  when taking into the account the expected manufacturing variations in the dielectric coating  1318  and airgaps  1340  in addition to other variations in the coupler geometry and materials. 
   Although  FIG. 13  shows a dual stripline embodiment, alternatives such as a microstrip embodiment, a coplanar embodiment, or an embodiment without a reference plane may be used, as discussed above. Furthermore, conductors  1330 A and  1330 B may be a different width than conductors  1336 A and  1336 B. Also, dielectric  1312  may be different thickness than dielectric  1320 . 
     FIG. 14  shows a view in the plane orthogonal to that of FIG.  13 . The flexible circuit  1350  for daughter card  1355  is folded into a circular loop, with the longitudinal axis of the signal conductors  1330 A and  1330 B (see  FIG. 13 ) lying along the loop circumference. The ends of the conductive signal traces  1330 A and  1330 B are connected to conductive etches on the two outer faces of the daughter card  1355  in order to provide connection to the transceiver and terminating resistors mounted on the daughter card  1355 . 
   The loop is then pressed onto the top surface of the motherboard  1365  so that the longitudinal axes of each motherboard conductor  1336 A and  1336 B is parallel with, and in the desired proximity to, the corresponding coupled flex circuit conductor to form an electromagnetic coupler  1360 . The length of the flexible circuit and vertical position of the daughter card are adjusted by mechanical means such that the motherboard conductors are in the desired proximity to the flex circuit conductors for a length L, which is selected to ensure that the capacitive and inductive coupling coefficients fall within the desired range of values. The length L may be 1 cm for example. 
   Some bandwidth reduction may be present in the flex strip implementation of  FIG. 14  if the flex strip is made of polyimide (dielectric constant=3.8) and the motherboard is made of FR4 glass-epoxy (dielectric constant=4.5). These materials are commercially available from well-known vendors such as 3M or DuPont. This may be eliminated if the FR4 is replaced with a material with a dielectric constant equal or close to that of polyimide, like Rogers RO4003 or similar lower dielectric constant materials. Rogers RO4003 is available from the Rogers Corporation. In the embodiment where the coupler is buried in the motherboard, the bandwidth may be limited by the dielectric losses in the FR4 material used in low-cost PCB assemblies. Again, the use of materials with lower dielectric losses like Rogers RO4003 relieves these limits. 
     FIG. 15  shows a detail of the contact area between the flexible circuit and the top surface of the motherboard corresponding to the embodiment outlined in  FIGS. 13 and 14 . Arranging the motherboard conductors  1336 A,  1336 B, in selected proximity to the flex circuit conductors  1330 A,  1330 B optionally having airgaps  1340  in between, creates the coupler. The motherboard-connected segments are lying in a plane where adjacent segments are arranged with an alternating angular displacement about the longitudinal axis of the conductor. The flex circuit conductors, similarly segmented, are arranged so that the angular displacements of its segments are in opposite sense to the corresponding segments in the motherboard. The composite structure may thus have the zig-zag geometry as shown in FIG.  6 A. 
   These and other embodiments of the present invention may be realized in accordance with these teachings and it should be evident that various modifications and changes may be made in these teachings without departing from the broader spirit and scope of the invention. The specification and drawings are, accordingly, to be regarded in an illustrative rather than restrictive sense and the invention measured only in terms of the claims.