Patent Publication Number: US-9407014-B2

Title: Antenna device

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     The present application claims priority to Japanese Patent Application No. 2012-240832 filed on Oct. 31, 2012, and to Japanese Patent Application No. 2013-218903 filed on Oct. 22, 2013, the entire contents of each of these applications being incorporated herein by reference in their entirety. 
     TECHNICAL FIELD 
     The technical field relates to an antenna device that includes a plurality of radiation elements and where isolation between radiation elements is enhanced. 
     BACKGROUND 
     Owing to a MIMO (Multi-Input Multi-Output) transmission technology where a plurality of radiation elements are installed on both of a transmitting side and a receiving side and spatial multiplexing is performed, it may be possible to perform high-speed and high-capacity wireless communication. Lowering of coupling and lowering of a correlation between a plurality of radiation elements are desired for a MIMO antenna. In Japanese Unexamined Patent Application Publication No. 2011-109440, Japanese Unexamined Patent Application Publication No. 2011-205316, Japanese Unexamined Patent Application Publication No. 2009-521898 (Translation of PCT Application), or Japanese Unexamined Patent Application Publication No. 2010-525680 (Translation of PCT Application), a technique has been disclosed where coupling between antenna elements is reduced by connecting two antenna elements to each other using a connection element. In the technique disclosed in Japanese Unexamined Patent Application Publication No. 2011-109440, as a connection element used for lowering of coupling, a variable reactance circuit is used. 
     SUMMARY 
     The present disclosure provides an antenna device capable of easily shifting an operating frequency band in a state where isolation between two radiation elements is maintained at a high level. 
     According to an embodiment of the present disclosure, an antenna device includes a first radiation element, a second radiation element, a first port configured to feed power to the first radiation element, a second port configured to feed power to the second radiation element, and a decoupling circuit configured to connect the first radiation element and the second radiation element. The decoupling circuit includes a bridge element connecting a first point between the first port and the first radiation element and a second point between the second port and the second radiation element to each other, a first reactance element provided in series with the first radiation element between the first point and the first radiation element, and a second reactance element provided in series with the second radiation element between the second point and the second radiation element, and at least one of the first reactance element and the second reactance element is capable of changing a value of reactance. 
     In a more specific embodiment, each of the first radiation element and the second radiation element may also be configured so as to resonate in a first frequency band and a second frequency band higher than the first frequency band. 
     In another more specific embodiment, a configuration is adopted where a first matching circuit inserted between the first port and the first point and a second matching circuit inserted between the second port and the second point are included. Each of the first matching circuit and the second matching circuit may be configured so as to achieve impedance matching in the first frequency band and the second frequency band. 
     Other features, elements, characteristics and advantages will become more apparent from the following detailed description with reference to the attached drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  is an equivalent circuit diagram of an antenna device according to a first embodiment, and  FIG. 1B  is a schematic perspective view of the antenna device according to the first exemplary embodiment. 
         FIG. 2  is a graph illustrating simulation results of S (Scattering) parameters in an initial state, a first state, and a second state of the antenna device according to the first embodiment. 
         FIG. 3  is a graph illustrating simulation results of S (Scattering) parameters of an antenna device according to a comparative example. 
         FIG. 4A  is an equivalent circuit diagram of an antenna device according to a second exemplary embodiment, and 
         FIG. 4B  is a schematic perspective view of the antenna device according to the second embodiment. 
         FIG. 5  is a graph illustrating simulation results of S (Scattering) parameters in a fifth state and a sixth state of the antenna device according to the second embodiment. 
     
    
    
     DETAILED DESCRIPTION 
     The inventor realized that in a technique of the related art, it has been difficult to shift an operating frequency band. A technique according to the present disclosure that shifts a frequency band in which an antenna device operates will now be described. 
       FIG. 1A  illustrates the equivalent circuit diagram of an antenna device according to a first exemplary embodiment. Power is fed from the first port  10  of the antenna device to a first radiation element  11 , and power is fed from the second port  20  thereof to a second radiation element  21 . The first radiation element  11  and the second radiation element  21  are configured so as to resonate at a single resonant frequency. The first port  10  and the second port  20  are connected to a transmission and reception circuit  30 . The transmission and reception circuit  30  is compatible with, for example, an MIMO transmission system. A decoupling circuit  40  connects the first port  10 , the second port  20 , the first radiation element  11 , and the second radiation element  21  to one another. 
     The decoupling circuit  40  includes a bridge element  41 , a first reactance element  12 , and a second reactance element  22 . The bridge element  41  connects a first point  13  between the first port  10  and the first radiation element  11  and a second point  23  between the second port  20  and the second radiation element  21  to each other. The first reactance element  12  is inserted in series with the first radiation element  11  between the first point  13  and the first radiation element  11 . The second reactance element  22  is inserted in series with the second radiation element  21  between the second point  23  and the second radiation element  21 . 
     At least one of the first reactance element  12  and the second reactance element  22  is configured so as to be capable of changing the value of reactance. As an example, variable inductors or variable capacitors are used for the first reactance element  12  and the second reactance element  22 . In addition, in each of the first reactance element  12  and the second reactance element  22 , a plurality of fixed inductors may also be disposed whose inductances are different, and one fixed inductor may also be selected using a switch. A fixed inductor or a fixed capacitor is used for the bridge element  41 . 
     A first matching circuit  14  is inserted between the first port  10  and the first point  13 , and a second matching circuit  24  is inserted between the second port  20  and the second point  23 . 
     A return loss when power is fed from the first port  10  to the first radiation element  11  is expressed as S 11 , and a transmission coefficient with the second port  20  is expressed as S 21 . In addition, a return loss when power is fed from the second port  20  to the second radiation element  21  is expressed as S 22 , and a transmission coefficient with the first port  10  is expressed as S 12 . The decoupling circuit  40  reduces the transmission coefficients S 21  and S 12 . In other words, isolation between the first radiation element  11  and the second radiation element  21  is enhanced. 
       FIG. 1B  illustrates the schematic perspective view of the antenna device according to the first embodiment. In the vicinity of the edge of a ground plate  50  having a substantially planar shape of a substantially rectangle shape, a high-frequency circuit  51  is disposed. The high-frequency circuit  51  includes the decoupling circuit  40 , the first matching circuit  14 , the second matching circuit  24  ( FIG. 1A ), and transmission lines connecting these circuits. The transmission line is configured using, for example, a microstrip line. A reactance element and a capacitance element, included in the high-frequency circuit  51 , are configured using lumped parameter elements or distributed constant circuits. 
     For example, planar monopole antennas are used for the first radiation element  11  and the second radiation element  21 . The first radiation element  11  and the second radiation element  21  are disposed in a slightly outer side portion of one side of the ground plate  50 . One end of each of the first radiation element  11  and the second radiation element  21  is connected to the high-frequency circuit  51 . 
     As a substrate for forming the ground plate  50 , a dielectric plate such as, for example, a glass epoxy resin can be used. For example, an ABS resin can be used for a carrier for forming the first radiation element  11  and the second radiation element  21 . In  FIG. 1B , no dielectric plate and no carrier are illustrated. 
     The S parameters of the antenna device according to the first embodiment were calculated owing to simulation. As a condition for the simulation, it was assumed that a dimension Y 1  in the vertical direction of the ground plate  50  illustrated in  FIG. 1B  and a dimension X in the lateral direction thereof were about 100 mm and about 60 mm, respectively, and the thickness of the ground plate  50  was about 1 mm. It was assumed that a dimension Y 2  in the vertical direction of an antenna region in which the first radiation element  11  and the second radiation element  21  are disposed and a dimension X in the lateral direction thereof were about 10 mm and about 60 mm, respectively. The first radiation element  11  and the second radiation element  21  have geometric forms substantially plane-symmetrical with respect to each other. The length of each of the first radiation element  11  and the second radiation element  21  is about 27.4 mm, and a distance between the two is about 5.2 mm. Copper was used for the ground plate  50 , the first radiation element  11 , and the second radiation element  21 . 
     Each of the first radiation element  11  and the second radiation element  21  is configured so as to resonate at a single resonant frequency of about  850  MHz. The decoupling circuit  40  illustrated in  FIG. 1A  was configured so that the transmission coefficient S 21  becomes a local minimum at a frequency of about 850 MHz. Specifically, inductors L 1  and L 2  are used for the first reactance element  12  and the second reactance element  22 , respectively, and both the inductances thereof are about 3.28 nH. An inductor LB is also used for the bridge element  41 , and the inductance thereof is about 3.52 nH. 
     The first matching circuit  14  and the second matching circuit  24  were configured so that the return losses S 11  and S 22  become local minimums at a frequency of about 850 MHz. Specifically, the first matching circuit  14  and the second matching circuit  24  were configured using shunt inductances of about 6.5 nH and series capacitances of about 5.0 pF. It is assumed that the above-mentioned state is referred to as an initial state Q 0 . 
     Under the condition that the transmission coefficients S 21  and S 12  represent local minimum values at a frequency of about 750 MHz lower than about 850 MHz, the element constants of the first reactance element  12  and the second reactance element  22  were calculated. At this time, the circuit constants of the bridge element  41 , the first matching circuit  14 , and the second matching circuit  24  are not changed. Under the above-mentioned condition, the inductances of the first reactance element  12  and the second reactance element  22  were about 6.10 nH. It is assumed that this state is referred to as a first state Q 1 . 
     In the same way, under the condition that the transmission coefficients S 21  and S 12  represent local minimum values at a frequency of about 950 MHz higher than about 850 MHz, the element constants of the first reactance element  12  and the second reactance element  22  were calculated. As a result, the inductances of the first reactance element  12  and the second reactance element  22  were about 1.25 nH. It is assumed that this state is referred to as a second state Q 2 . 
       FIG. 2  illustrates simulation results of S (Scattering) parameters of the antenna device at the times of the initial state Q 0 , the first state Q 1 , and the second state Q 2 . In a horizontal axis, a frequency is expressed in unit of “GHz”, and in a vertical axis, the magnitudes of S (Scattering) parameters are expressed in unit of “dB”. Solid lines illustrated in  FIG. 2  indicate the transmission coefficient S 21 , and dashed lines indicate the return loss S 11 . The thickest lines indicate the initial state Q 0 , the second thickest lines indicate the first state Q 1 , and the thinnest lines indicate the second state Q 2 . In the initial state Q 0 , both of the transmission coefficient S 21  and the return loss S 11  represent local minimum values at a frequency of about  850  MHz, according to design targets. In addition, owing to the substantial symmetry of radiation elements and circuits, the return loss S 22  is approximately equal to the return loss S 11 , and the transmission coefficient S 12  is approximately equal to the transmission coefficient S 21 . 
     In the first state Q 1 , the transmission coefficient S 21  represents a local minimum value at about 750 MHz, according to a design target. At this time, the return loss S 11  also represents a local minimum value at about 750 MHz. Therefore, at the time of the first state Q 1 , it may be possible for the antenna device to efficiently operate in a frequency band located near a frequency of about 750 MHz. 
     In the second state Q 2 , the transmission coefficient S 21  represents a local minimum value at about 950 MHz, according to a design target. At this time, the return loss S 11  also represents a local minimum value at about 950 MHz. Therefore, at the time of the second state Q 2 , it may be possible for the antenna device to efficiently operate in a frequency band located near a frequency of about 950 MHz. 
     With reference to  FIG. 3 , the simulation results of the S (Scattering) parameters of an antenna device according to a comparative example will be described. In the comparative example, the inductances of the inductors L 1  and L 2  in the first reactance element  12  and the second reactance element  22  illustrated in  FIG. 1A  were fixed, and the circuit constant of the bridge element  41  was changed. The initial state Q 0  of the antenna device in the comparative example is approximately the same as the initial state Q 0  of the antenna device ( FIG. 1A ,  FIG. 1B , and  FIG. 2 ) according to the first embodiment. 
     When the circuit constant of the bridge element  41  was calculated under the condition that the transmission coefficients S 21  and S 12  represent local minimum values at about 750 MHz (the first state Q 1 ), the inductance of the bridge element  41  was about 13.0 nH. When the circuit constant of the bridge element  41  was calculated under the condition that the transmission coefficients S 21  and S 12  represent local minimum values at about 950 MHz (the second state Q 2 ), the bridge element  41  changed to a capacitive property, and the capacitance thereof was about 27 pF. 
       FIG. 3  illustrates the simulation results of the S (Scattering) parameters of the antenna device according to the comparative example at the times of the initial state Q 0 , the first state Q 1 , and the second state Q 2 . In a horizontal axis, a frequency is expressed in unit of “GHz”, and in a vertical axis, the magnitudes of S (Scattering) parameters are expressed in unit of “dB”. Solid lines illustrated in  FIG. 3  indicate the transmission coefficient S 21 , and dashed lines indicate the return loss S 11 . The thickest line indicates the initial state Q 0 , the second thickest line indicates the first state Q 1 , and the thinnest line indicates the second state Q 2 . 
     In the first state Q 1 , the transmission coefficient S 21  represents a local minimum value at about 750 MHz, according to a design target. However, the return loss S 11  represents a local minimum value at about 780 MHz, and is deviated away from a frequency at which the transmission coefficient S 21  takes a local minimum value. Since the return loss S 11  is large at about 750 MHz, the antenna device according to the comparative example is not suitable for an operation in a frequency band located near about 750 MHz. 
     In the second state Q 2 , the transmission coefficient S 21  represents a local minimum value at about 950 MHz, according to a design target. In addition, the return loss S 11  also represents a local minimum value at about 950 MHz. However, compared with the S (Scattering) parameters in the second state Q 2  in  FIG. 2 , it is understood that valleys are shallow that occur at about 950 MHz in the second state Q 2  of the antenna device according to the comparative example. In other words, compared with the second state Q 2  of the antenna device according to the first embodiment illustrated in  FIG. 2 , isolation between the first port  10  and the second port  20  ( FIG. 1A ) is weak, and the return loss S 11  is large. Therefore, the antenna device according to the comparative example is not suitable for an operation in a frequency band located near about 950 MHz. 
     As described above, in the antenna device according to the first embodiment, the circuit constant of the bridge element  41  illustrated in  FIG. 1A  is fixed, and the circuit constants of the first reactance element  12  and the second reactance element  22  are made variable. Owing to this, it may become possible to shift a frequency band in which the antenna device operates, and after the operating frequency band has been shifted, it may be possible to maintain the small transmission coefficient S 21  (high isolation) and the small return loss S 11 . 
     Furthermore, in the first embodiment, in order to change the operating frequency band from about 750 MHz to about 950 MHz, it is only necessary to change the inductances of the inductors L 1  and L 2  from about 1.25 nH to about 6.10 nH. The amount of change therein is about 4.85 nH. On the other hand, in the comparative example illustrated in  FIG. 3 , in order to change the operating frequency band from about 850 MHz to about 750 MHz, it is necessary to change the inductance of the bridge element  41  from about 3.52 nH to about 13.0 nH, and the amount of change therein is about 9.48 nH. Furthermore, in the comparative example, in order to change the operating frequency band from about 850 MHz to about 950 MHz, it is necessary to change the bridge element  41  from an induction property to a capacitive property. In this way, in the first embodiment, compared with the comparative example, it may be possible to reduce the amount of change in a circuit constant, which is used for shifting the operating frequency band. 
     With reference to  FIG. 4A ,  FIG. 4B , and  FIG. 5 , an antenna device according to a second exemplary embodiment will be described. Hereinafter, a difference from the first embodiment will be described, and the description of the same configuration described above will not be repeated. In the first embodiment, the first radiation element  11  and the second radiation element  21  ( FIG. 1B ) are configured so as to resonate at a single resonant frequency. In the second embodiment, the first radiation element  11  and the second radiation element  21  are configured so as to resonate at two resonant frequencies. As an example, in the first radiation element  11  and the second radiation element  21 , two-resonance characteristics are obtained using a fundamental and a harmonic. 
       FIG. 4A  illustrates the equivalent circuit diagram of the antenna device according to the second embodiment. In the second embodiment, as described above, the first radiation element  11  and the second radiation element  21  have two-resonance characteristics. Variable capacitors CB 1  and CB 2  are used for the first reactance element  12  and the second reactance element  22 , respectively. A T-type circuit is used for the first matching circuit  14 , and the first matching circuit  14  includes a series inductor LD 1 , a shunt inductor LC 1 , and a series capacitor CC 1 . As an example, the inductance of the series inductor LD 1  is about 1.5 nH, the inductance of the shunt inductor LC 1  is about 9 nH, and the capacitance of the series capacitor CC 1  is about 5 pF. The second matching circuit  24  also has the same configuration, and includes a series inductor LD 2 , a shunt inductor LC 2 , and a series capacitor CC 2 . An inductor LB is used for the bridge element  41 , and the inductance thereof is about 4 nH. 
       FIG. 4B  illustrates the schematic perspective view of the antenna device according to the second embodiment. As the first radiation element  11  and the second radiation element  21 , inverted-F antennas are used. The transmission and reception circuit  30  ( FIG. 4A ) feeds power to the feeding points of the first radiation element  11  and the second radiation element  21  through the high-frequency circuit  51 . 
     The circuit constants of the first reactance element  12  and the second reactance element  22  were changed, and the S (Scattering) parameters of the antenna device were calculated owing to simulation. It is assumed that a state where the capacitances of the variable capacitors CB 1  and CB 2  in the first reactance element  12  and the second reactance element  22  are set to about 8 pF is referred to as a third state Q 3  and a state where the capacitances of the variable capacitors CB 1  and CB 2  are set to about 1 pF is referred to as a fourth state Q 4 . 
       FIG. 5  illustrates the simulation results of the transmission coefficient S 21  and the return loss S 11  when the antenna device according to the second embodiment is in the third state Q 3  and the fourth state Q 4 . In a horizontal axis, a frequency is expressed in unit of “GHz”, and in a vertical axis, the magnitudes of S (Scattering) parameters are expressed in unit of “dB”. Solid lines in  FIG. 5  indicate the transmission coefficient S 21 , and dashed lines indicate the return loss S 11 . Thick lines indicate the third state Q 3 , and thin lines indicate the fourth state Q 4 . 
     When the antenna device is in the third state Q 3 , the transmission coefficient S 21  and the return loss S 11  represent local minimum values in a first frequency band  61 A located near about 700 MHz. Furthermore, in a second frequency band  62 A located near about 1.75 GHz, the transmission coefficient S 21  and the return loss S 11  represent local minimum values in a second frequency band  62 A located near about 1.75 GHz. Therefore, when being in the third state Q 3 , the antenna device may efficiently operate in both of the first frequency band  61 A and the second frequency band  62 A. 
     When the antenna device is in the fourth state Q 4 , the transmission coefficient S 21  and the return loss S 11  represent local minimum values in a first frequency band  61 B located near about 880 MHz. Furthermore, in a second frequency band  62 B located near about 2 GHz, the transmission coefficient S 21  and the return loss S 11  represent local minimum values. Therefore, when being in the fourth state Q 4 , the antenna device may efficiently operate in both of the first frequency band  61 B and the second frequency band  62 B. 
     In the second embodiment, in the same way as the second embodiment, it may also be possible to shift operating frequency bands on both of the low-frequency wave side and the high-frequency wave side. Even if the operating frequency bands are shifted, it may be possible to maintain high isolation and a low return loss. 
     The first matching circuit  14  and the second matching circuit  24  are designed so as to achieve impedance matching in the first frequency bands  61 A and  61 B and the second frequency bands  62 A and  62 B. 
     With embodiments according to the present disclosure, by changing the value of the reactance of at least one of the first reactance element and the second reactance element, it is possible to shift a frequency at which a transmission coefficient between the first port and the second port becomes a local minimum. After the shift of the frequency, it is also possible to maintain a small return loss. 
     While exemplary embodiments have been described above, it is to be understood that variations, modifications, improvements, and combinations may occur without departing from the scope and spirit of the disclosure.