Patent Publication Number: US-2016226148-A1

Title: Laminated waveguide, wireless module, and wireless system

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2015-020431, filed on Feb. 4, 2015, the entire contents of which are incorporated herein by reference. 
     FIELD 
     The embodiments discussed herein are related to a laminated waveguide, a wireless module, and a wireless system. 
     BACKGROUND 
     Conventionally, there has been a planar antenna module comprising an antenna section, a feedline section, and a connected conductor. The antenna section includes a first ground conductor having a first slot, a second ground conductor having a dielectric, an antenna substrate having a radiating element, a third ground conductor having a dielectric, and a fourth ground conductor. The feedline section includes the fourth ground conductor, a fifth ground conductor, a feed substrate, a sixth ground conductor, and a seventh ground conductor, and the connected conductor includes a second waveguide opening. The planar antenna module is formed by laminating a connection conductor with a high frequency circuit, the seventh ground conductor, the sixth ground conductor, the feed substrate, the fifth ground conductor, the fourth ground conductor, the third ground conductor, the antenna substrate, the second ground conductor, and the first ground conductor in this order (See International Publication Pamphlet No. WO 2006/098054). 
     Incidentally, since a conventional planar antenna module has a complicated structure and desires high-precision positioning when the planar antenna module is assembled, there is a problem that manufacturing cost is expensive. 
     Hence, an objective is to provide a laminated waveguide, a wireless module, and a wireless system that reduce the manufacturing cost. 
     SUMMARY 
     According to an aspect of the invention, a laminated waveguide includes: a dielectric layer, a first and a second patch antennae formed on a first face of the dielectric layer, a third and a fourth patch antennae formed on a second face of the dielectric layer, the first face being opposite to the second surface, a first and a second transmission lines formed on the dielectric layer and connected to the first and the second patch antennae, respectively, a third and a fourth transmission lines formed on the dielectric layer and connected to the third and the fourth patch antennae, respectively, wherein a pair of the first and the third patch antennae and another pair of the second and the fourth patch antennae are arranged to form an angle between the pair and the another pair so as to suppress interference between the pair and the another pair. 
     The object and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the claims. 
     It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention, as claimed. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIGS. 1A and 1B  are diagrams illustrating a wireless communication module including a laminated waveguide of an embodiment 1, and a wireless communication system; 
         FIG. 2  is a perspective view illustrating the laminated waveguide of the embodiment 1; 
         FIG. 3  is a view illustrating an exploded state of the laminated waveguide illustrated in  FIG. 2 ; 
         FIG. 4  is a top view illustrating the laminated waveguide; 
         FIG. 5  is a view illustrating a cross section taken along the arrow V-V in  FIG. 4 ; 
         FIGS. 6A and 6B  are views illustrating a simulation model of the laminated waveguide; 
         FIGS. 7A, 7B, and 7C  are diagrams illustrating simulation results of S parameters and bandwidths; 
         FIGS. 8A and 8B  are views illustrating distribution of an electric field in a model used in the simulation; 
         FIGS. 9A, 9B, and 9C  are diagrams illustrating dependence of a resonant frequency, the S parameters, and bandwidths on a combination of length and diameter; 
         FIG. 10  is a perspective view illustrating a laminated waveguide of an embodiment 2; 
         FIG. 11  is a view illustrating an exploded state of the laminated waveguide illustrated in  FIG. 10 ; 
         FIG. 12  is a top view illustrating the laminated waveguide; 
         FIG. 13  is a view illustrating a cross section taken along the arrow XIII-XIII in  FIG. 12 ; 
         FIGS. 14A and 14B  are views illustrating a simulation model of the laminated waveguide; 
         FIG. 15  is a characteristic diagram illustrating a relationship between dimensions of a patch antenna and input impedance; 
         FIG. 16  is a diagram summarizing in a tabular format simulation results obtained when width is changed; 
         FIG. 17  is a graph illustrating frequency characteristics of S 11  parameter, S 21  parameter, S 41  parameter, and S 42  parameter in the laminated waveguide; 
         FIGS. 18A and 18B  are views illustrating a simulation model of a laminated waveguide according to a variant of the embodiment 2; 
         FIG. 19  is a diagram summarizing in a tabular format simulation results when the width is changed in the laminated waveguide of the variant of the embodiment 2; 
         FIG. 20  is a graph illustrating frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter when the width is set to 0.49 mm in the laminated waveguide of the variant of the embodiment 2; and 
         FIG. 21  is a view illustrating a configuration of a laminated waveguide according to the variant of the embodiment 2. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Embodiments to which a laminated waveguide, a wireless module, and a wireless system of the present disclosure are applied are described hereinafter. 
     Embodiment 1 
       FIGS. 1A and 1B  are diagrams illustrating a wireless communication module  50  including a laminated waveguide  100  of embodiment 1, and a wireless communication system  500 .  FIG. 1A  is a block diagram and  FIG. 1B  is a side view illustrating an example of an installation state. 
     As illustrated in  FIG. 1A , the wireless communication system  500  has an antenna  510 , the wireless communication module  50 , and a baseband signal processing section  520 . 
     The wireless communication module  50  includes the laminated waveguide  100 , a monolithic microwave integrated circuit (MMIC) module  51 , and an MMIC drive circuit  52 . 
     The MMIC module  51  is a device connected to the laminated waveguide  100  and configured to perform wireless front-end processing. In the MMIC module  51 , an amplifier, a mixer, an oscillator (voltage-controlled oscillator: VCO), a multiplexer and the like are integrated. The MMIC module  51  is configured to generate a high-frequency signal of a millimeter waveband (hereinafter referred to as a millimeter wave) transmitted from the antenna  510  and extract a difference between a reflection signal received by the antenna  510  and a transmitted high-frequency signal. 
     The MMIC drive circuit  52  is a circuit configured to drive the MMIC module  51 . 
     The baseband signal processing section  520  processes a low-frequency component corresponding to the difference in frequencies to retrieve desired information. The baseband signal processing section  520  is an example of the signal processing section. 
     The laminated waveguide  100  of the wireless communication module  50  has a simple configuration, low transmission loss, and good isolation characteristics, which thus allows downsizing of the wireless communication module  50  and cost reduction to be achieved. 
     In addition, using another different wireless communication system  500 , the wireless communication system  500  may perform communications in millimeter waves between the two wireless communication systems  500 . The communications in millimeter waves may narrow directivity, thus easily enabling multiple channels. 
     In addition, the wireless communication system  500  may be used as a radar device. A distance to an object may be measured based on a time difference between a radio wave that the wireless communication system  500  emits from the antenna  510  and a radio wave that is reflected and then received. In addition, if the laminated waveguide  100  has waveguides for multiple channels and the wireless communication system  500  includes antennae  510  for the multiple channels, a distance to an object is measured by the multiple antennae  510  aligned in parallel, thereby allowing a direction in which the object is located to be detected based on a difference in the distance. 
     In addition, as illustrated in  FIG. 1B , in the wireless communication system  500 , by way of example, the antenna  510  is mounted on one surface  100 A of the laminated waveguide  100 , and the MMIC module  51 , the MMIC drive circuit  52 , and the baseband signal processing section  520  are mounted on other surface  100 B. 
     The laminated waveguide  100  includes patch antennae  160 A,  170 A and transmission lines  180 A,  190 A. The patch antenna  160 A and the transmission line  180 A are formed on the surface  100 A. The patch antenna  160 A is connected to the antenna  510  through the transmission line  180 A. 
     The patch antenna  170 A and the transmission line  190 A are formed on the surface  100 B. The patch antenna  170 A is connected to the MMIC module  51  through the transmission line  190 A. The MMIC module  51  is connected to the MMIC drive circuit  52  through a wiring layer  53  formed on the surface  100 B, and the MMIC drive circuit  52  is connected to the baseband signal processing section  520  through a wiring layer  54  formed on the surface  100 B. 
     Since the patch antennae  160 A and  170 A build a waveguide, the antenna  510  and the MMIC module  51  are connected through the waveguide built by the patch antennae  160 A and  170 A. 
     Now, the antenna  510  is mounted on the opposite side of the laminated waveguide  100  to the MMIC module  51  and the MMIC drive circuit  52  because it is intended not to allow the antenna  510  to receive a high-frequency signal generated by the MMIC module  51  and the MMIC drive circuit  52 . 
     For example, even if the transmission lines  180 A and  190 A are connected by using a general wiring substrate instead of the laminated waveguide  100  and a contact plug or the like instead of the patch antennae  160 A and  170 A, it is still difficult to transmit a millimeter wave through the contact plug or the like. 
     For these reasons, the laminated waveguide  100  wherein the patch antenna  160 A on the side of the surface  100 A and the patch antenna  170 A on the side of the surface  100 B build the waveguide is used. 
     A configuration of the laminated waveguide  100  is described hereinafter. 
       FIG. 2  is a perspective view illustrating the laminated waveguide  100  of the embodiment 1.  FIG. 3  is a view illustrating an exploded state of the laminated waveguide  100  illustrated in  FIG. 2 .  FIG. 4  is a top view illustrating the laminated waveguide  100 .  FIG. 5  is a view illustrating a cross section taken along the arrow V-V in  FIG. 4 . Note that an XYZ coordinate system (orthogonal coordinate system) is defined below, as illustrated in  FIGS. 2 to 5 . 
     The laminated waveguide  100  includes a dielectric layer  110 , a conductive layer  120 , a dielectric layer  130 , a conductive layer  140 , a dielectric layer  150 , patch antennae  160 A,  160 B,  170 A,  170 B, and transmission lines  180 A,  180 B,  190 A,  190 B. 
     Here, a mode in which the dielectric layer  110 , the conductive layer  120 , the dielectric layer  130 , the conductive layer  140 , the dielectric layer  150 , the patch antennae  160 A,  160 B,  170 A,  170 B, and the transmission lines  180 A,  180 B,  190 A,  190 B are achieved by a wiring substrate of the flame retardant 4 (FR4) standard is described. 
       FIGS. 2 to 5  illustrate a part of the dielectric layer  110 , the conductive layer  120 , the dielectric layer  130 , the conductive layer  140 , and the dielectric layer  150  to be achieved by the wiring substrate. More specifically, the dielectric layer  110 , the conductive layer  120 , the dielectric layer  130 , the conductive layer  140 , and the dielectric layer  150  actually extends more in the X-axis direction and the Y-axis direction than a rectangular part illustrated in a plan view in  FIGS. 2 to 5 . 
     The dielectric layer  110  is made of a dielectric (insulator). The dielectric layer  110  is an example of a first dielectric layer. As the dielectric layer  110 , a core material, fiberglass is impregnated with epoxy resin fiberglass, may be used. The conductive layers  120  and  140  are formed on both sides of the dielectric layer  110  as the core material. 
     The conductive layer  120  is provided on a surface on the side in the positive Z-axis direction of the dielectric layer  110 . The conductive layer  120  is an example of a first conductive layer. The conductive layer  120  may be made of metal such as copper or aluminum or the like. As described above, when a core material is used as the dielectric layer  110 , copper foil or the like attached to one side of the core material (surface on the side in the positive Z-axis direction) may be used as the conductive layer  120 . 
     The conductive layer  120  has slots  121 A,  121 B. The slots  121 A,  121 B are respectively an example of a first slot and a second slot. The slots  121 A,  121 B are circular in a plan view and openings penetrating through the conductive layer  120  in the thickness direction (Z-axis direction). Diameters of the slots  121 A,  121 B are equal to each other. 
     The slots  121 A,  121 B may be formed by patterning the copper foil attached to the surface on the side in the positive Z-axis direction of the core material as the dielectric layer  110 , for example, with the photolithography method and the wet etching method. 
     By way of example, the slots  121 A and  121 B are located such that the center point of width of the conductive layer  120  in the Y-axis direction coincides with the center point between the openings and such that the slots are symmetrical with respect to a line, as an axis of symmetry, passing in the Y-axis direction through the center point of the width of the conductive layer  120  in the X-axis direction. 
     The dielectric layer  130  is made of a dielectric (insulator) and laminated in the positive Z-axis direction of the conductive layer  120 . The dielectric layer  130  is an example of a second dielectric layer. As described above, when a core material is used as the dielectric layer  110 , a prepreg layer, in which fiberglass is impregnated with epoxy resin, for example, may be used as the dielectric layer  130 . 
     The conductive layer  140  is provided on a surface on the side in the negative Z-axis direction of the dielectric layer  110 . The conductive layer  140  is an example of a second conductive layer. The conductive layer  140  may be made of metal such as copper or aluminum or the like. As described above, when a core material is used as the dielectric layer  110 , copper foil attached to the surface on the side in the negative Z-axis direction of the core material may be used as the conductive layer  140 . 
     The conductive layer  140  has slots  141 A,  141 B. The slots  141 A,  141 B are respectively an example of a third slot and a fourth slot. The slots  141 A,  141 B are circular in a plan view and openings penetrating through the conductive layer  140  in the thickness direction (Z-axis direction). Diameters of the slots  141 A,  141 B are equal to diameters of the slots  121 A,  121 B formed on the conductive layer  120 . 
     The slots  141 A,  141 B may be formed by patterning the copper foil attached to the surface on the side in the negative Z-axis direction of the core material as the dielectric layer  110 , for example, with the photolithography method and the wet etching method. 
     By way of example, the slots  141 A and  141 B are located such that the center point of width of the conductive layer  120  in the Y-axis direction coincides with the center point between the openings and such that the slots are symmetrical with respect to a line, as an axis of symmetry, passing in the Y-axis direction through the center point of the width of the conductive layer  120  in the X-axis direction. 
     More specifically, the slots  141 A,  141 B are aligned respectively with the slots  121 A,  121 B in a plan view. In other words, the slots  141 A,  141 B are respectively formed at positions corresponding to the slots  121 A,  121 B in a plan view. 
     The dielectric layer  150  is made of a dielectric (insulator), similarly to the dielectric layer  130 , and laminated in the negative Z-axis direction of the conductive layer  140 . The dielectric layer  150  is an example of a third dielectric layer. As described above, when a core material is used as the dielectric layer  110 , a prepreg layer, in which fiberglass is impregnated with epoxy resin, for example, may be used as the dielectric layer  150 . 
     The patch antennae  160 A,  160 B are respectively located within the slots  121 A,  121 B on the surface on the side in the positive Z-axis direction of the dielectric layer  130 , in a plan view. The patch antennae  160 A,  160 B are respectively an example of a first patch antenna and a second patch antenna. The patch antennae  160 A,  160 B may be made of metal such as copper or aluminum or the like. 
     As described above, when a core material is used as the dielectric layer  110 , the patch antennae  160 A,  160 B may be formed by patterning copper foil attached to a surface on the side in the positive Z-axis direction of the dielectric layer  130 , for example, with the photolithography method and the wet etching method. 
     The patch antenna  160 A is oblong (rectangle) in a plan view, and a length of the longitudinal direction is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency. The patch antenna  160 A is formed such that angle θ 1  which a center axis L 1  parallel to the longitudinal direction forms to the X-axis is 45 degrees. 
     Similarly, the patch antenna  160 B is oblong (rectangle) in a plan view and a length of the longitudinal direction is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency. The patch antenna  160 B is formed such that angle θ 2  which a center axis L 2  parallel to the longitudinal direction forms to the X-axis is 45 degrees. 
     The angle θ 1  is an angle in a direction rotating anti-clockwise with respect to the X-axis and the angle θ 2  is an angle in a direction rotating clockwise with respect to the X-axis. Thus, the patch antennae  160 A and  160 B are in a positional relation in which the center axes L 1 , L 2  are rotated 45 degrees in opposite directions to each other, with respect to the X-axis. 
     In addition, the transmission line  180 A is connected to an end section  161 A of the patch antenna  160 A in the longitudinal direction. Since the end section  161 A is located on the center axis L 1 , the end section  161 A is located at the center between edge sides of the patch antenna  160 A in the short direction (direction orthogonal to the longitudinal direction in a plan view). Here, an end section on the opposite side to the end section  161 A in the longitudinal direction is an end section  162 A. 
     Since the patch antenna  160 A has a configuration described above, the end section  161 A serves as a power feed point when power is fed to the patch antenna  160 A from the transmission line  180 A. In addition, an electric field in the end sections  161 A and  162 A is largest at this moment, and an electric field at a mid-point between the end sections  161 A and  162 A is zero. 
     More specifically, the patch antenna  160 A emits in the Z-axis direction a sine wave-like radio wave amplitude of which changes in a direction of the center axis L 1  connecting the end sections  161 A and  162 A. 
     In addition, the transmission line  180 B is connected to an end section  161 B of the patch antenna  160 B in the longitudinal direction. Since the end section  161 B is located on the center axis L 2 , the end section  161 B is located at the center edge sides of the patch antenna  160 B in the short direction (direction orthogonal to the longitudinal direction in a plan view). Here, an end section on the opposite side to the end section  161 B in the longitudinal direction is an end section  162 B. 
     The patch antenna  160 B is similar to the patch antenna  160 A except that the angle θ 2  to the X-axis is different from the angle θ 1  of the patch antenna  160 A. Thus, the end section  161 B serves as a power feed point when power is fed to the patch antenna  160 B from the transmission line  180 B. In addition, an electric field in the end sections  161 B and  162 B is largest at this moment, and an electric field at a mid-point between the end sections  161 B and  162 B is zero. 
     More specifically, the patch antenna  160 B emits in the Z-axis direction a radio wave amplitude of which changes in a direction of the center axis L 2  connecting the end sections  161 B and  162 B. 
     When power is fed to the patch antennae  160 A,  160 B as described above, an electric field Em in directions in which the center axes L 1 , L 2  extend is generated on the patch antennae  160 A,  160 B. 
     The patch antennae  170 A,  170 B are respectively located within the slots  141 A,  141 B on the surface on the side in the negative Z-axis direction of the dielectric layer  150 , in a plan view. The patch antennae  170 A,  170 B are respectively an example of a third patch antenna and a fourth patch antenna. The patch antennae  170 A,  170 B may be made of metal such as copper or aluminum or the like. 
     As described above, when a core material is used as the dielectric layer  110 , the patch antennae  170 A,  170 B may be formed by patterning copper foil attached to a surface on the side in the negative Z-axis direction of the dielectric layer  150 , for example, with the photolithography method and the wet etching method. 
     The patch antenna  170 A is oblong (rectangle) in a plan view, and a length of the longitudinal direction is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency. The patch antenna  170 A is equal in size to the patch antenna  160 A in a plan view and a position on an XY plane is equal to that of the patch antenna  160 A. More specifically, the patch antenna  170 A coincides with the patch antenna  160 A in a plan view. 
     Thus, the patch antenna  170 A is formed such that an angle which a center axis parallel to the longitudinal direction (center axis coinciding with the center axis L 1  in a plan view) forms to the X-axis is 45 degrees. 
     Similarly, the patch antenna  170 B is oblong (rectangle) in a plan view, and a length of the longitudinal direction is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency. The patch antenna  170 B is equal in size to the patch antenna  160 B in a plan view and a position on an XY plane is equal to that of the patch antenna  160 B. More specifically, the patch antenna  170 B coincides with the patch antenna  160 B in a plan view. 
     Thus, the patch antenna  170 B is formed such that an angle which a center axis parallel to the longitudinal direction (center axis coinciding with the center axis L 2  in a plan view) forms to the X-axis is 45 degrees. 
     More specifically, the patch antennae  170 A and  170 B are in a positional relation in which the center axes parallel to the longitudinal directions are rotated 45 degrees in opposite directions to each other, with respect to the X-axis. 
     Here, both end sections of the patch antenna  170 A in the longitudinal direction, which are at the same positions in a plan view as the end sections  161 A,  162 A of the patch antenna  160 A, are end sections  171 A,  172 A. Similarly, both end sections of the patch antenna  170 B in the longitudinal direction, which are at the same positions in a plan view as the end sections  161 B,  162 B of the patch antenna  160 B, are end sections  171 B,  172 B. 
     The transmission line  190 A is connected to the end section  172 A of the patch antenna  170 A in the longitudinal direction (See  FIG. 3 ). Since the end section  172 A is located on a center axis parallel to the longitudinal direction, the end section  172 A is located at the center between edge sides of the patch antenna  170 A in the short direction (direction orthogonal to the longitudinal direction in a plan view). 
     Since the patch antenna  170 A has a configuration as described above, the end section  172 A serves as a power feed point when power is fed to the patch antenna  170 A from the transmission line  190 A. In addition, an electric field in the end sections  171 A and  172 A is largest at this moment, and an electric field at a mid-point between the end sections  171 A and  172 A is zero. 
     More specifically, the patch antenna  170 A emits in the Z-axis direction a sine wave-like radio wave amplitude of which changes in a direction of the center axis connecting the end sections  171 A and  172 A. Accordingly, the patch antenna  170 A can communicate with the patch antenna  160 A. In addition, positioning the patch antennae  160 A and  170 A so that the patch antennae  160 A and  170 A correspond to each other at the same angle and fit within the slots  121 A,  141 A makes communications in a radiated electromagnetic field efficient and easy to perform. 
     In addition, the transmission line  190 B is connected to the end section  172 B of the patch antenna  170 B in the longitudinal direction. Since the end section  172 B is located on a center axis parallel to the longitudinal direction, the end section  172 B is located at the center between edge sides of the patch antenna  170 B in the short direction (direction orthogonal to the longitudinal direction in a plan view). 
     The end section  172 B serves as a power feed point when power is fed to the patch antenna  170 B from the transmission line  190 B. In addition, an electric field in the end sections  171 B and  172 B is largest at this moment, and an electric field at a mid-point between the end sections  171 B and  172 B is zero. 
     More specifically, the patch antenna  170 B emits in the Z-axis direction a radio wave amplitude of which changes in a direction of the center axis connecting the end sections  171 B and  172 B. Accordingly, the patch antenna  170 B can communicate with the patch antenna  160 B. In addition, positioning the patch antennae  160 B and  170 B so that the patch antennae  160 B and  170 B correspond to each other at the same angle and fit within the slots  121 B,  141 B makes communications in a radiated electromagnetic field efficient and easy to perform. One ends of the transmission lines  180 A,  180 B are respectively connected to the end sections  161 A,  161 B of the patch antennae  160 A,  160 B. In addition, other ends of the transmission lines  180 A,  180 B are connected to an antenna device or an integrated circuit, or the like. The transmission lines  180 A,  180 B are respectively an example of a first transmission line and a second transmission line. Note that the antenna device or the integrated circuit or the like connected to the other ends of the transmission lines  180 A,  180 B are omitted in  FIGS. 2 to 5 . 
     The transmission lines  180 A,  180 B are laminated on the conductive layer  120  with the dielectric layer  130  in between and build a microstrip line with the conductive layer  120 . Characteristic impedance of the transmission lines  180 A,  180 B is set to 50Ω, by way of example. A length between one ends and other ends of the transmission lines  180 A,  180 B is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency of the patch antennae  160 A,  160 B. 
     One ends of the transmission lines  190 A,  190 B are respectively connected to the end sections  172 A,  172 B of the patch antennae  170 A,  170 B. In addition, other ends of the transmission lines  190 A,  190 B are connected to a circuit that generates a high-frequency signal, or the like. The transmission lines  190 A,  190 B are respectively an example of a third transmission line and a fourth transmission line. Note that the circuit that is connected to the other ends of the transmission lines  190 A,  190 B and generates a high-frequency signal, or the like is omitted in  FIGS. 2 to 5 . 
     The transmission lines  190 A,  190 B are laminated on the conductive layer  140  with the dielectric layer  150  in between and build a microstrip line with the conductive layer  140 . Characteristic impedance of the transmission lines  190 A,  190 B is set to 50Ω, by way of example. A length between one ends and other ends of the transmission lines  190 A,  190 B is set to an electrical length that is half of a wavelength λ (λ/2) in a resonant frequency of the patch antennae  170 A,  170 B. 
     In the laminated waveguide  100  having the configuration as described above, a direction of amplitude of a radio wave emitted by the patch antenna  160 A in the Z-axis direction is the direction which forms the angle θ 1  (45 degrees) in an anticlockwise direction to the X-axis in a plan view, and a direction of amplitude of a radio wave emitted by the patch antenna  160 B in the Z-axis direction is the direction which forms the angle θ 2  (45 degrees) in a clockwise direction to the X-axis in a plan view. 
     In addition, the direction of the amplitude of the radio wave emitted by the patch antenna  170 A in the Z-axis direction is equal to the direction of the amplitude of the radio wave emitted by the patch antenna  160 A in the Z-axis direction, and the direction of the amplitude of the radio wave emitted by the patch antenna  170 B in the Z-axis direction is equal to the direction of the amplitude of the radio wave emitted by the patch antenna  160 B in the Z-axis direction. 
     Thus, an angle made by the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A and the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B is 90 degrees. More specifically, the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A is orthogonal to the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B. 
     Here, as described above, the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A is orthogonal to the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B. 
     Therefore, even when a radio wave emitted by the patch antennae  160 A and  170 A leaks to the patch antennae  160 B and  170 B, it is possible to avoid reception of the radio wave by the patch antennae  160 B and  170 B. 
     Conversely, even when a ratio wave emitted by the patch antennae  160 B and  170 B leaks to the patch antennae  160 A and  170 A, it is possible to avoid reception of the radio wave by the patch antennae  160 A and  170 A. 
     More specifically, the patch antennae  160 A and  170 A, and the patch antennae  160 B and  170 B being positioned such that amplitude directions of electric fields are orthogonal to each other in a plan view, isolation of a waveguide built by the patch antennae  160 A and  170 A from a waveguide built by the patch antennae  160 B and  170 B is improved. 
     With such a configuration, interference of a radio wave being transmitted through the waveguide of the patch antennae  160 A and  170 A with a radio wave being transmitted through the waveguide of the patch antennae  160 B and  170 B is suppressed. 
     Simulation results are described hereinafter with reference to  FIGS. 6A to 9C . 
       FIGS. 6A and 6B  are views illustrating a simulation model of the laminated waveguide  100 .  FIGS. 7A, 7B, and 7C  are diagrams illustrating simulation results of S parameters and bandwidths.  FIGS. 8A and 8B  are views illustrating distribution of an electric field in a model used in the simulation. 
     As illustrated in  FIG. 6A , a diameter of the slots  121 A,  121 B,  141 A,  141 B is Sr, a distance between the centers of the slots  121 A and  121 B is PD, and line width of the transmission lines  180 A,  180 B,  190 A,  190 B is W. Note that angles θ 1 , θ 2  are the same as those illustrated in  FIG. 4 . 
     In addition, as illustrated in  FIG. 6B , a length of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction is PL, and a length in the short direction is PS. 
     First, an optimal value of the length PS of the patch antennae  160 A,  160 B,  170 A,  170 B in the short direction is determined. Since a value of input impedance Z 11  which is close to 50Ω is obtained when the length PS is 0.4 mm, it is now decided to perform simulation, fixing the length PS to 0.4 mm. 
     The length PL of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction was set to 1.0 mm, the length PS in the short direction to 0.4 mm, a thickness to 0.1 mm, and the diameter Sr of the slots  121 A,  121 B,  141 A,  141 B to 1.35 mm. In addition, a line length of the transmission lines  180 A,  180 B,  190 A,  190 B was set to λ/4 when a resonant frequency Fc was 78.0 GHz, line width W of the transmission lines  180 A,  180 B,  190 A,  190 B to 0.16 mm, and distance PD between the centers of the slots  121 A,  121 B to 2.0 mm. Note that a thickness of the transmission lines  180 A,  180 B,  190 A,  190 B is 0.1 mm. 
     In addition, a thickness of the dielectric layer  110  was set to 1 mm and a relative permittivity of the dielectric layer  110  to 3.8, a thickness of the dielectric layers  130  and  150  to 0.14 mm and a relative permittivity of the dielectric layers  130  and  150  to 4.4, and a thickness of copper foil used as the conductive layers  120  and  140  to 0.1 mm. 
     Here, to determine S 11  parameter, S 21  parameter, S 41  parameter, and S 42  parameter, the transmission line  190 A was assigned to Port 1 , the transmission line  180 A to Port 2 , the transmission line  190 B to Port 3 , and the transmission line  180 B to Port 4 . 
     In addition, as a model of a laminated waveguide for comparison, a model with both angles θ 1 , θ 2  set to 0 degrees was used (See  FIG. 8A ). 
       FIG. 7A  is a graph illustrating frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide for comparison.  FIG. 7B  is a graph illustrating frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide  100 .  FIG. 7C  is a diagram summarizing contents of  FIGS. 7A and 7B  in a tabular format. 
     Here, as a bandwidth BW 1 , a band whose value of the S 11  parameter was less than −10 dB was evaluated. As a bandwidth BW 2 , a band whose value of the S 21  parameter was higher than −6 dB was evaluated. In addition, as a bandwidth BW 4 , a band whose values of the S 41  parameter and the S 42  parameters were both less than −22 dB was evaluated. 
     In the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide for comparison as illustrated in  FIG. 7A , the bandwidths BW 1 , BW 2 , BW 4  were 8.0 GHz, 7.0 GHz, and 0.2 GHz, respectively. 
     Since the value of BW 4  is small, in particular, it is seen that signals are transmitted between Port 1  and Port 4  and between Port 2  and Port 4 . In other words, it is seen that Port 4  interferes with a transmission line between Port 1  and Port 2 . 
       FIG. 8A  illustrates distribution of electric fields in the model of the laminated waveguide for comparison, and  FIG. 8B  is a view illustrating distribution of electric fields in the model of the laminated waveguide  100 . As illustrated in  FIG. 8A  in the model of the laminated waveguide for comparison in which both angles θ 1 , θ 2  are set to 0 degrees, the patch antennae  160 A and  160 B are parallel to the X-axis. 
     In  FIG. 8A , an area with a larger electric field is illustrated in darker gray, and an area with a smaller electric field is illustrated in white or light gray. 
     As illustrated in  FIG. 8A , in the model of the laminated waveguide for comparison, the strong electric field illustrated in dark gray is also generated at Port 4  when a signal is conducted from Port 1  to Port 2 , from which it is seen that Port 4  interferes with the transmission line between Port 1  and Port 2 . 
     As such, in the model of the laminated waveguide for comparison, it was learned that Port 4  interfered with the transmission line between Port 1  and Port 2 . 
     In contrast to this, in the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide  100  as illustrated in  FIG. 7B , the bandwidths BW 1 , BW 2 , BW 4  were respectively 8.0 GHz, 2.0 GHz, and 3.7 GHz. 
     Since the value of BW 4  is improved, in particular, it is seen that interference of the transmission line between Port 1  and Port 2  with Port 4  is reduced and isolation is improved to some extent. 
     In addition, as illustrated in  FIG. 8B , in the model of the laminated waveguide  100 , no strong electric field expressed in dark gray is generated in Port 4  when a signal is conducted from Port 1  to Port 2 , from which it is seen that Port 4  is isolated from the transmission line between Port 1  and Port 2 . 
     As such, in the model of the laminated waveguide  100 , it was learned that interference of the transmission line between Port 1  and Port 2  with Port 4  was reduced and that isolation was improved to some extent. 
     The above results are as illustrated in  FIG. 7C . In the laminated waveguide for comparison with both angles θ 1 , θ 2  set to 0 degrees, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter when the resonant frequency Fc was 78.0 GHz were −2.5 dB, 18.6 dB, and −15.1 dB, respectively. 
     In addition, the bandwidths BW 1 , BW 2 , and BW 4  were 8.0 GHz, 7.0 GHz, and 0.2 GHz, respectively. A band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 0.2 GHz from 81.2 GHz to 81.4 GHz. 
     In contrast to this, in the laminated waveguide  100  with both angles θ 1 , θ 2  set to 45 degrees, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter when the resonant frequency Fc was 78.0 GHz were −2.9 dB, −24.9 dB, and −27.0 dB, respectively. 
     More specifically, it was learned that improvement of about 6 dB to about 12 dB was made compared with the laminated waveguide for comparison, and that isolation was improved. 
     In addition, the bandwidths BW 1 , BW 2 , and BW 4  were 8.0 GHz, 6.0 GHz, and 3.7 GHz, respectively. A band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 3.7 GHz from 75.0 GHz to 78.7 GHz. 
       FIGS. 9A, 9B, and 9C  are diagrams illustrating dependence of the resonant frequency Fc, the S parameters, BW 1 , BW 2 , BW 4 , and BW on a combination of the length PL and the diameter Sr. 
     As illustrated in  FIG. 9A , the following was learned by changing the length PL of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction and the diameter Sr of the slots  121 A,  121 B,  141 A,  141 B. 
     When the diameter Sr was increased to 1.08 mm, 1.22 mm, and 1.35 mm with the length PL fixed to 1.0 mm, as illustrated in  FIG. 9B , the resonant frequency Fc decreased and was 78.2 GHz when the diameter Sr was 1.35 mm. 
     Good values were obtained all for the S 21  parameter, the S 41  parameter, and the S 42  parameter. 
     The value of BW 1  increased as the diameter Sr of the slots  121 A,  121 B,  141 A,  141 B increased. Leakage of radio waves increasing, the value of BW 1  increased. In contrast to this, there was not much change in the value of BW 2 . 
     In addition, as the diameter Sr of the slots  121 A,  121 B,  141 A,  141 B increased, the value of BW 4  decreased. It is considered that this is because radio waves leaking to the outside of the laminated waveguide  100  from the slots  121 A,  121 B,  141 A,  141 B increased as the diameter Sr increased. 
     Note that the dependence of the BW 1 , BW 2 , BW 3 , BW 4  on the diameter Sr is as illustrated in  FIG. 9C . 
     In addition, when the length PL was increased to 1.0 mm, 1.1 mm, and 1.2 mm with the diameter Sr fixed to 1.35 mm, the resonant frequency Fc decreased. 
     Relatively good values were obtained for the S 21  parameter, the S 41  parameter, and the S 42  parameter. 
     As described above, in the embodiment 1, the so-called wiring substrate structure was utilized to fabricate the laminated waveguide  100  which includes the waveguide built by the patch antennae  160 A and  170 A and the waveguide built by the patch antennae  160 B and  170 B. 
     Therefore, according to the embodiment 1, the laminated waveguide  100 , the wireless communication module  50 , and the wireless communication system  500  with the manufacturing cost reduced may be provided. 
     In addition, in the laminated waveguide  100  of the embodiment 1, as described above, the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A was made orthogonal to the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B. 
     Therefore, the laminated waveguide  100  may be provided wherein isolation of the waveguide built by the patch antennae  160 A and  170 A from the waveguide built by the patch antennae  160 B and  170 B is improved and interference between radio waves transmitted in the waveguides is reduced. 
     Note that a mode is not limited to the mode in which the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A is orthogonal to the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B. 
     It is learned from a simulation trend that interference between waveguides is considerably reduced and isolation is improved if angle formed by the direction of the amplitude of the radio wave emitted by the patch antennae  160 A and  170 A and the direction of the amplitude of the radio wave emitted by the patch antennae  160 B and  170 B is about 90 degrees±15 degrees. 
     In addition, for example, when the antenna  510  is mounted on one surface  100 A and the MMIC module  51  and the MMIC drive circuit  52  are mounted on the other surface  100 B, as illustrated in  FIG. 1B , the laminated waveguide  100  is very effective, since a high-frequency signal generated by the MMIC module  51  and the MMIC drive circuit  52  is not easily received by the antenna  510 . 
     In addition, in the above, the mode that the laminated waveguide  100  includes waveguides for two channels built by the patch antennae  160 A,  160 B,  170 A,  170 B and the slots  121 A,  121 B,  141 A,  141 B is described. 
     However, the laminated waveguide  100  may be in such a configuration that the laminated waveguide  100  has three or more waveguides by including more patch antennae and slots. 
     Embodiment 2 
       FIG. 10  is a perspective view illustrating a laminated waveguide  200  of an embodiment 2.  FIG. 11  is a view illustrating an exploded state of the laminated waveguide  200  illustrated in  FIG. 10 .  FIG. 12  is a top view illustrating the laminated waveguide  200 .  FIG. 13  is a view illustrating a cross section taken along the arrow XIII-XIII in  FIG. 12 . Note that an XYZ coordinate system (orthogonal coordinate system) is defined hereinafter, as illustrated in  FIGS. 10 to 13 . 
     The laminated waveguide  200  of the embodiment 2 has a configuration to which bridges are added that divide into two the slots  121 A,  121 B,  141 A,  141 B, respectively, of the laminated waveguide  100  of the embodiment 1. Thus, components similar to the components of the laminated waveguide  100  are assigned the same symbols and a description the components is omitted. 
     The laminated waveguide  200  includes a dielectric layer  110 , a conductive layer  220 , a dielectric layer  130 , a conductive layer  240 , a dielectric layer  150 , patch antennae  160 A,  160 B,  170 A,  170 B, and transmission lines  180 A,  180 B,  190 A,  190 B. 
     Here, a mode in which the dielectric layer  110 , the conductive layer  220 , the dielectric layer  130 , the conductive layer  240 , the dielectric layer  150 , the patch antenna  160 A,  160 B,  170 A,  170 B, and the transmission lines  180 A,  180 B,  190 A,  190 B are achieved by a wiring substrate of the FR4 standard is described. 
     The conductive layers  220  and  240  are formed on both sides of the dielectric layer  110 . 
     The conductive layer  220  has slots  221 A,  221 B and bridges  222 A,  222 B. There are two each for the slots  221 A and  221 B. The bridges  222 A and  222 B are both examples of a first bridge. 
     The bridge  222 A crosses between the two slots  221 A, and the bridge  222 B crosses between the two slots  221 B. The slots  221 A,  221 B have a configuration in which the slots  121 A,  121 B of the embodiment 1 are respectively divided into two by the bridges  222 A,  222 B. 
     The bridge  222 A is arranged such that the bridge  222 A passes through the center of a virtual circle made by the two slots  221 A and is orthogonal to the center axis L 1 . The virtual circle is equal to the opening of the slot  121 A of the embodiment 1. 
     The bridge  222 B is arranged such that the bridge  222 B passes through the center of a virtual circle made by the two slots  221 B and is orthogonal to the center axis L 1 . The virtual circle is equal to the opening of the slot  121 B of the embodiment 1. 
     Thus, as illustrated in  FIG. 12 , the bridges  222 A,  222 B are respectively orthogonal to the patch antennae  160 A,  160 B (in the longitudinal direction) in a plan view. 
     The slots  221 A,  221 B which are respectively divided into two by such bridges  222 A,  222 B can be formed by patterning copper foil attached to a surface on the side in the positive Z-axis direction of a core material as the dielectric layer  110 , for example, with the photolithography method and the wet etching method. 
     The conductive layer  240  has slots  241 A,  241 B and bridges  242 A,  242 B. There are two each for the slots  241 A,  241 B. The bridges  242 A,  242 B are both examples of a second bridge. 
     The bridge  242 A crosses between the two slots  241 A, and the bridge  242 B crosses between the two slots  241 B. The slots  241 A,  241 B have a configuration in which the slots  141 A,  141 B of the embodiment 1 are respectively divided into two by the bridges  242 A,  242 B. 
     The bridge  242 A is arranged such that the bridge  242 A passes through the center of a virtual circle made by the two slots  241 A and is orthogonal to the center axis which is parallel to the longitudinal direction of the patch antenna  170 A. The virtual circle is equal to the opening of the slot  141 A of the embodiment 1. 
     The bridge  242 B is arranged such that the bridge  242 B passes through the center of a virtual circle made by the two slots  241 B and is orthogonal to the center axis which is parallel to the longitudinal direction of the patch antenna  170 B. The virtual circle is equal to the opening of the slot  141 B of the embodiment 1. 
     Thus, the bridges  242 A,  242 B are respectively orthogonal to the patch antennae  170 A,  170 B (in the longitudinal direction) in a plan view. 
     The slots  241 A,  241 B are respectively aligned with the slots  221 A,  221 B in a plan view. In other words, the slots  241 A,  241 B are formed at positions corresponding to the slots  221 A,  221 B in a plan view. Thus, the bridges  242 A,  242 B are respectively aligned with the bridges  222 A,  222 B, in a plan view. 
     The slots  241 A,  241 B which are respectively divided into two by such bridges  242 A,  242 B can be formed by patterning the copper foil attached to the surface on the side in the positive Z-axis direction of a core material as the dielectric layer  110 , for example, with the photolithography method and the wet etching method. 
     The bridges  222 A,  222 B extend in a direction orthogonal to a direction (direction in which the center axes L 1 , L 2  extend) in which an amplitude of a radio wave emitted by the patch antennae  160 A,  160 B in the Z-axis direction fluctuates. 
     Thus, electric potentials of the bridges  222 A,  222 B are fixed in the direction in which the bridges  222 A,  222 B extend. 
     Therefore, even when an electric field Es leaks in the width direction of the patch antennae  160 A,  160 B, fluctuations of the electric field Es leaking in the width direction of the patch antennae  160 A,  160 B are limited by the bridges  222 A,  222 B. Thus, in the patch antennae  160 A,  160 B, the electric field Em in the direction in which the center axes L 1 , L 2  extend is dominantly generated. 
     In addition, the bridge  222 A passes through the center of the virtual circle formed by the two slots  241 A, and the bridge  222 B passes through the center of the virtual circle formed by the two slots  241 B. Thus, the bridges  222 A,  222 B respectively pass through the center points of the patch antennae  160 A,  160 B in the longitudinal direction. Therefore, electric potentials of the bridges  222 A,  222 B are 0 V in the direction in which the bridges  222 A,  222 B extend. 
     Therefore, the bridges  222 A,  222 B, respectively, hardly limit an amplitude of an electric field which is generated in the longitudinal direction of the patch antennae  160 A,  160 B. 
     In addition, a relation between the bridges  242 A,  242 B and the patch antennae  170 A,  170 B is similar to a relation between the bridges  222 A,  222 B and the patch antennae  160 A,  160 B. 
     Thus, even when an electric field leaks in the width direction of the patch antennae  170 A,  170 B, fluctuations of the electric field Es leaking in the width direction of the patch antennae  170 A,  170 B are limited by the bridges  242 A,  242 B. 
     In addition, the bridges  242 A,  242 B, respectively, hardly limit an amplitude of an electric field which is generated in the longitudinal direction of the patch antennae  170 A,  170 B. 
     As described above, according to the embodiment 2, the patch antennae  160 A and  170 A and the patch antennae  160 B and  170 B being positioned such that amplitude directions of electric fields are orthogonal to each other in a plan view, isolation of a waveguide built by the patch antennae  160 A and  170 A from a waveguide built by the patch antennae  160 B and  170 B can be improved. This is similar to the laminated waveguide  100  of the embodiment 1. 
     In addition, according to the embodiment 2, in addition to the effect described above, even when an electric field leaks in the width direction of the patch antennae  160 A,  160 B,  170 A,  170 B, fluctuations in the electric field leaking in the width direction are limited by the bridges  222 A,  222 B,  242 A,  242 B. 
     The bridges  222 A,  222 B,  242 A,  242 B, respectively, hardly limit an amplitude of an electric field which is generated in the longitudinal direction of the patch antennae  160 A,  160 B,  170 A,  170 B. 
     Therefore, according to the embodiment 2, the laminated waveguide  200  can be provided in which the isolation of a waveguide built by the patch antennae  160 A and  170 A from a waveguide built by the patch antennae  160 B and  170 B is further improved. 
     Simulation results are described hereinafter with reference to  FIGS. 14A to 17 . 
       FIGS. 14A and 14B  are views illustrating a simulation model of the laminated waveguide  200 .  FIG. 15  is a characteristic diagram illustrating a relationship between dimensions of a patch antenna  160 A and input impedance Z 11 . As illustrated in  FIG. 14A , diameter of the slots  221 A,  221 B,  241 A,  241 B is Sr, a distance between the centers of the slots  221 A and  221 B is PD, and line width of the transmission lines  180 A,  180 B,  190 A,  190 B is W. This is similar to  FIG. 6B . In addition, width of the bridges  222 A,  222 B,  242 A,  242 B is Sh. Note that angles θ 1 , θ 2  are the same as those illustrated in  FIG. 4 . 
     In addition, as illustrated in  FIG. 14B , a length of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction is PL, and a length in the short direction is PS. This is similar to  FIG. 6B . 
     First, a length of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction PL was set to 1.0 mm, a thickness of the patch antennae  160 A,  160 B,  170 A,  170 B was set to 0.1 mm, and a length in the short direction PS was changed. Then, characteristics of input impedance Z 11  as illustrated in  FIG. 15  were obtained. 
     As illustrated in  FIG. 15 , when the length in the short direction PS was changed from 0.4 mm to 0.9 mm, the input impedance Z 11  changed from approximately 65Ω to approximately 108Ω. Since a value closest to 50Ω was obtained when the length PS was 0.4 mm, it was now decided to perform simulation, fixing the length PS to 0.4 mm. 
     Here, the length PL of the patch antennae  160 A,  160 B,  170 A,  170 B in the longitudinal direction was set to 1.0 mm, the length PS in the short direction to 0.4 mm, a thickness to 0.1 mm, and the diameter Sr of the slots  221 A,  221 B,  241 A,  241 B to 1.35 mm. In addition, line length of the transmission lines  180 A,  180 B,  190 A,  190 B was set to λ/4 when a resonant frequency Fc was 78.0 GHz, line width W of the transmission lines  180 A,  180 B,  190 A,  190 B to 0.03 mm, and distance PD between the centers of the slots  221 A,  221 B to 2.0 mm. Note that a thickness of the transmission lines  180 A,  180 B,  190 A,  190 B is 0.1 mm. 
     In addition, a thickness of the dielectric layer  110  was set to 1 mm and a relative permittivity of the dielectric layer  110  to 3.8, a thickness of the dielectric layers  130  and  150  to 0.14 mm and a relative permittivity of the dielectric layers  130  and  150  to 4.4, and a thickness of copper foil used as the conductive layers  120  and  140  to 0.1 mm. 
     First, when the width Sh of the bridges  222 A,  222 B,  242 A,  242 B was changed, results illustrated in  FIG. 16  were obtained. 
       FIG. 16  is a diagram summarizing in a tabular format simulation results obtained when width Sh is changed. 
     Here, a band a value of S 11  parameter of which was less than −10 dB as a bandwidth BW 1  was evaluated. As a bandwidth BW 2 , a band a value of the S 21  parameter of which was higher than −6 dB was evaluated. In addition, as a bandwidth BW 4 , a band values of S 41  parameter and S 42  parameter of which were both less than −22 dB was evaluated. 
     In the laminated waveguide  200 , a simulation was performed by setting the width Sh to 0 mm, 0.37 mm, and 0.49 mm. Note that the width Sh being 0 mm is a case in which the bridges  222 A,  222 B,  242 A,  242 B are not present, and the case corresponds to the configuration of the laminated waveguide  100  of the embodiment 1. 
     When the width Sh is 0 mm, at a point at which the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −2.9 dB, −24.9 dB, and −27.0 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 8.0 GHz, 6.0 GHz, and 3.8 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 3.8 GHz from 75.0 GHz to 78.8 GHz. 
     When the width Sh was 0.37 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −4.6 dB, −31.4 dB, and −27.0 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 7.7 GHz, 7.6 GHz, and 5.6 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 5.6 GHz from 75.2 GHz to 80.6 GHz. 
     When the width Sh is 0.49 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −3.1 dB, −34.5 dB, and −25.2 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 6.2 GHz, 6.4 GHz, and 3.8 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 3.8 GHz from 75.0 GHz to 78.8 GHz. 
     As described above, when the width Sh was 0.37 mm, the BW exhibited the best value of 5.6. The BWs of when the width Sh was 0 mm and when the width Sh was 0.49 mm were both 3.8. 
     With this, it was learned that in order to obtain the good isolation characteristics, it is important to set the width Sh of the bridges  222 A,  222 B,  242 A,  242 B to appropriate width which is not too thick. 
     Then, S parameters were determined by setting the width Sh of the bridges  222 A,  222 B,  242 A,  242 B to 0.37 mm, as illustrated in  FIG. 17 . 
       FIG. 17  is a graph illustrating the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide  200 . 
     In the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter in the laminated waveguide  200  as illustrated in  FIG. 17 , the bandwidths BW 1 , BW 2 , BW 4  were 7.7 GHz, 7.6 GHz, and 5.6 GHz, respectively. 
     Since the BW 2  and the BW 4  are improved, in particular, it is learned that interference of the transmission line between Port 1  and Port 2  with Port 4  is more reduced than in the laminated waveguide  100  of the embodiment 1, and the isolation is improved. 
     As such, in the model of the laminated waveguide  200 , it is learned that interference of the transmission line between Port 1  and Port 2  with Port 4  is more reduced than in the laminated waveguide  100  of the embodiment 1, and the isolation is improved. 
     As described above, according to the embodiment 2, the bridges  222 A,  222 B,  242 A,  242 B being provided, fluctuations in an electric field which leaks in the width direction are limited even when the electric field leaks in the width direction of the patch antennae  160 A,  160 B,  170 A,  170 B, and thus the good isolation characteristics can be obtained. 
     Therefore, according to the embodiment 2, the laminated waveguide  200  in which the isolation of a waveguide built by the patch antennae  160 A and  170 A from a waveguide built by the patch antennae  160 B and  170 B is further improved can be provided. 
     Note that in the above, the mode in which the bridges  222 A,  222 B,  242 A,  242 B are formed in each of the slots  221 A,  221 B,  241 A,  241 B is described. However, the bridges  222 A,  242 A may be respectively formed only in the slots  221 A,  241 A, and the bridges  222 B,  242 B may not be formed in the slots  221 B,  241 B. 
     Simulation results of a laminated waveguide according to a variant of the embodiment 2 are described hereinafter with reference to  FIGS. 18A to 20 . 
       FIGS. 18A and 18B  are views illustrating a simulation model of the laminated waveguide according to the variant of the embodiment 2. 
     The laminated waveguide of the variant of the embodiment 2 has the shape of the patch antennae  160 A,  160 B,  170 A,  170 B of the laminated waveguide  200  of the embodiment 2 that is changed from the rectangle to a regular hexagon. In the following, each component of the laminated waveguide of the variant is described using the same symbols as the respective components of the laminated waveguide  200  of the embodiment 2. 
     As illustrated in  FIG. 18A , diameter of slots  221 A,  221 B,  241 A,  241 B is Sr, a distance between the centers of the slots  221 A and  221 B is PD, and line width of the transmission lines  180 A,  180 B,  190 A,  190 B is W. In addition, width of bridges  222 A,  222 B,  242 A,  242 B is Sh. Note that angles θ|, θλ are the same as those illustrated in  FIG. 4 . 
     In addition, as illustrated in  FIG. 18B , a length in a direction in which an amplitude of an electric field of the hexagonal patch antennae  160 A,  160 B,  170 A,  170 B is generated is R 1 , and a length in a direction orthogonal to the length R 1  is R 2 . In addition, angle formed by four sides which are not orthogonal to the direction (direction of the length R 1 ) in which the amplitude of the electric field of the hexagonal patch antennae  160 A,  160 B,  170 A,  170 B is generated is θ 3 . 
     Here, the length R 1  was set to 1.0 mm, the length R 2  to 1.15 mm, the angle θ 3  to 30 degrees, a thickness to 0.1 mm, and diameter Sr of the slots  221 A,  221 B,  241 A,  241 BB to 1.35 mm. In addition, a line length of the transmission lines  180 A,  180 B,  190 A,  190 B was set to λ/4 of when a resonant frequency Fc was 78.0 GHz, line width W of the transmission lines  180 A,  180 B,  190 A,  190 B to 0.03 mm, and distance PD between the slots  221 A and  221 B to 2.0 mm. Note that a thickness of the transmission lines  180 A,  180 B,  190 A,  190 B is 0.1 mm. 
     In addition, a thickness of the dielectric layer  110  was set to 1 mm and a relative permittivity of the dielectric layer  110  to 3.8, a thickness of the dielectric layers  130  and  150  to 0.14 mm and a relative permittivity the dielectric layers  130  and  150  to 4.4, and a thickness of copper foil used as the conductive layers  120  and  140  to 0.1 mm. 
     First, when width Sh of the bridges  222 A,  222 B,  242 A,  242 B was changed to 0.0 mm, 0.37 mm, 0.49 mm, and 0.61 mm, results as illustrated in  FIG. 19  were obtained. 
     Note that the width Sh being 0 mm is a case in which the bridges  222 A,  222 B,  242 A,  242 B are not present. 
       FIG. 19  is a diagram summarizing in a tabular format simulation results when the width Sh was changed in the laminated waveguide of the variant of the embodiment 2. 
     Here, as a bandwidth BW 1 , a band whose value of S 11  parameter was less than −10 dB was evaluated. As a bandwidth BW 2 , a band whose value of S 21  parameter was higher than −6 dB was evaluated. In addition, as a bandwidth BW 4 , a band whose values of S 41  parameter and S 42  parameters were both less than −22 dB was evaluated. 
     When the width Sh was 0 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −5.3 dB, −22.2 dB, and −28.8 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 3.8 GHz, 2.1 GHz, and 1.4 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 1.4 GHz from 77.0 GHz to 78.4 GHz. 
     When the width Sh was 0.37 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −3.2 dB, −27.7 dB, and −25.6 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 8.0 GHz, 4.4 GHz, and 2.4 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 2.4 GHz from 76.4 GHz to 78.8 GHz. 
     When the width Sh was 0.49 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −3.2 dB, −32.6 dB, and −27.6 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 8.0 GHz, 4.8 GHz, and 5.4 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 4.4 GHz from 75.8 GHz to 79.6 GHz. 
     When the width Sh was 0.61 mm, at a point where the resonant frequency Fc was 78.0 GHz, the values of the S 21  parameter, the S 41  parameter, and the S 42  parameter were −3.9 dB, −37.8 dB, and −23.2 dB, respectively. In addition, the BW 1 , the BW 2 , and the BW 4  were 6.5 GHz, 4.6 GHz, and 3.4 GHz, respectively. Thus, a band BW in which all of the bandwidths BW 1 , BW 2 , and BW 4  indicated a better value than the evaluation standard mentioned above was 3.4 GHz from 76.0 GHz to 79.4 GHz. 
     As described above, when the width Sh was 0.49 mm, the BW exhibited the best value of 4.4. 
     With this, it is learned that in order to obtain the good isolation characteristics, it is important to set the width Sh of the bridges  222 A,  222 B,  242 A,  242 B to appropriate width which is not too thick. 
       FIG. 20  is a graph illustrating the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter when the width Sh is set to 0.49 mm in the laminated waveguide of the variant of the embodiment 2. 
     Here, as a bandwidth BW 1 , a band whose value of S 11  parameter was less than −10 dB was evaluated. As a bandwidth BW 2 , a band whose value of S 21  parameter was higher than −6 dB was evaluated. In addition, as a bandwidth BW 4 , a band whose values of S 41  parameter and S 42  parameter were both less than −22 dB was evaluated. 
     In the frequency characteristics of the S 11  parameter, the S 21  parameter, the S 41  parameter, and the S 42  parameter illustrated in  FIG. 20 , the bandwidths BW 1 , BW 2 , BW 4  were 8.0 GHz or higher (BW 1 ≧8.0), 4.8 GHz, and 5.4 GHz, respectively. 
     Since the BW 2  and the BW 4  are improved, in particular, it is learned that interference of the transmission line between Port 1  and Port 2  with Port 4  is more reduced than in the laminated waveguide  100  of the embodiment 1, and the isolation is improved. 
     As such, it is learned in the model of the laminated waveguide of the variant of the embodiment 2 that interference of the transmission line between Port 1  and Port 2  with Port 4  is more reduced than in the laminated waveguide  100  of the embodiment 1 and isolation is further improved. 
     As described above, according the variant of the embodiment 2, the hexagonal bridges  222 A,  222 B,  242 A,  242 B being provided, fluctuations of an electric field leaking in the width direction are limited even when the electric field leaks in the width direction of the patch antennae  160 A,  160 B,  170 A,  170 B, and good isolation characteristics can be obtained. 
     Therefore, according to the variant of the embodiment 2, the laminated waveguide can be provided in which the isolation of a waveguide built by the hexagonal patch antennae  160 A and  170 A from a waveguide built by the hexagonal patch antennae  160 B and  170 B is further improved. 
       FIG. 21  is a view illustrating a configuration of a laminated waveguide  200 A according to the variant of the embodiment 2. 
     The laminated waveguide  200 A is the laminated waveguide  200  illustrated in  FIG. 10  to which shield pins  250  are added. 
     In  FIG. 21 , the number of shield pins  250  is three, the shield pins being arranged along the direction of the X-axis between waveguides for two channels that are built by patch antennae  160 A,  160 B,  170 A,  170 B and slots  221 A,  221 B,  241 A  241 B. The shield pins  250  penetrate through a dielectric layer  110 , a conductive layer  220 , a dielectric layer  130 , a conductive layer  240 , and a dielectric layer  150  in the Z-axis direction, and are connected to the conductive layer  120  and the conductive layer  140 . Both ends of the shield pins  250  appear on surfaces of the dielectric layers  130  and  150 . 
     Use of such shield pins  250  can improve the isolation characteristics between waveguides for different channels. 
     Note that both ends of the shield pins  250  may not appear on the dielectric layers  130  and  150 , and may be formed between the conductive layers  220  and  240 . 
     While the laminated waveguides, the wireless modules, and the wireless systems of illustrative embodiments of the present disclosure are described above, the present disclosure is not to be limited to the specifically disclosed embodiments, and various variations or modifications may be made without deviating from the claims. 
     All examples and conditional language recited herein are intended for pedagogical purposes to aid the reader in understanding the invention and the concepts contributed by the inventor to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a showing of the superiority and inferiority of the invention. Although the embodiments of the present invention have been described in detail, it should be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention.