Patent Publication Number: US-10784903-B2

Title: Phase control for carrier aggregation

Description:
CROSS-REFERENCE TO RELATED APPLICATION(S) 
     This application is a continuation of U.S. application Ser. No. 14/683,512 filed Apr. 10, 2015, entitled CIRCUITS AND METHODS RELATED TO RADIO-FREQUENCY RECEIVERS HAVING CARRIER AGGREGATION, which claims priority to and the benefit of the filing date of U.S. Provisional Application No. 61/978,808 filed Apr. 11, 2014, entitled CIRCUITS AND METHODS RELATED TO RADIO-FREQUENCY RECEIVERS HAVING CARRIER AGGREGATION, the benefits of the filing dates of which are hereby claimed and the disclosures of which are hereby expressly incorporated by reference herein in their entirety. 
    
    
     BACKGROUND 
     Field 
     The present disclosure relates to carrier aggregation in radio-frequency applications. 
     Description of the Related Art 
     In some radio-frequency (RF) applications, cellular carrier aggregation (CA) can involve two or more RF signals being processed through a common path. For example, carrier aggregation can involve use of a path for a plurality of bands having frequency ranges that are sufficiently separated. In such a configuration, simultaneous operation of more than one band can be achieved. 
     SUMMARY 
     In accordance with a number of implementations, the present disclosure relates to a carrier aggregation (CA) circuit that includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, with the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open-circuit impedance for the second frequency band. The CA circuit further includes a second path implemented between the second filter and the common node, with the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open-circuit impedance for the first frequency band. 
     In some embodiments, the first filter and the second filter can be parts of a diplexer. The diplexer can include an input port configured to receive a radio-frequency (RF) signal from an antenna. The common node can be configured to be coupled to an input of a low-noise amplifier (LNA). The LNA can be configured to amplify frequency bands of the received RF signal corresponding to the first frequency band and the second frequency band. The first frequency band can include, for example, a B3 band having a frequency range of 1.805 to 1.880 GHz, and the second frequency band can include, for example, a B1 band having a frequency range of 2.110 to 2.170 GHz. The second frequency band can further include a B4 band having a frequency range of 2.110 to 2.155 GHz. 
     In some embodiments, the first frequency band can include a B2 band having a frequency range of 1.930 to 1.990 GHz, and the second frequency band can include a B2 band having a frequency range of 2.110 to 2.155 GHz. The first frequency band can further include a B25 band having a frequency range of 1.930 to 1.995 GHz. In some embodiments, the first frequency band can include a B2 band having a frequency range of 1.930 to 1.990 GHz, and the second frequency band can include a B4 band having a frequency range of 2.110 to 2.155 GHz. 
     In some embodiments, the first path can include a first phase shifting circuit, and the second path can include a second phase shifting circuit. In some embodiments, the first phase shifting circuit can include, for example, two series capacitances and an inductive shunt path that couples a node between the two capacitances and a ground. The second phase shifting circuit can include two series capacitances and an inductive shunt path that couples a node between the two capacitances and a ground. In some embodiments, at least some of the capacitances and inductances of the first and second phase shifting circuits can be implemented as lumped elements. In some embodiments, at least some of the capacitances and inductances of the first and second phase shifting circuits can be implemented as distributed elements. 
     In some embodiments, the first phase shifting circuit can include two series inductances and a capacitive shunt path that couples a node between the two inductances and a ground. The second phase shifting circuit can include two series inductances and a capacitive shunt path that couples a node between the two inductances and a ground. In some embodiments, at least some of the capacitances and inductances of the first and second phase shifting circuits can be implemented as lumped elements. In some embodiments, at least some of the capacitances and inductances of the first and second phase shifting circuits can be implemented as distributed elements. 
     In some embodiments, each of the first path and the second path can include a switch to allow the CA circuit to operate in a CA mode or a non-CA mode. The switch for the first path can be at an output of the first phase shifting circuit and the switch for the second path can be at an output of the second phase shifting circuit. Both switches for the first and second paths can be closed for the CA mode. One of the two switches can be closed and the other switch can be closed for the non-CA mode. 
     In some implementations, the present disclosure relates to a radio-frequency (RF) module having a packaging substrate configured to receive a plurality of components, and a carrier aggregation (CA) circuit implemented on the packaging substrate. The CA circuit includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, with the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open-circuit impedance for the second frequency band. The CA circuit further includes a second path implemented between the second filter and the common node, with the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open-circuit impedance for the first frequency band. 
     In some embodiments, each of the first filter and the second filter can include a surface acoustic wave (SAW) filter. The first SAW filter and the second SAW filter can be implemented as a diplexer. In some embodiments, the RF module can further include a low-noise amplifier (LNA) implemented on the packaging substrate. The LNA can be coupled to the common node to receive a combined signal from the first path and the second path. In some embodiments, the RF module can be a front-end module. In some embodiments, the RF module can be a diversity receive (DRx) module. 
     In some embodiments, the first path can include a first phase shifting circuit, and the second path can include a second phase shifting circuit. Each of the first phase shifting circuit and the second phase shifting circuit can include capacitance and inductance elements. At least some of the capacitance and inductance elements can be implemented as passive devices mounted on or within the packaging substrate. 
     In a number of teachings, the present disclosure relates to a method for fabricating a radio-frequency (RF) module. The method includes providing or forming a packaging substrate configured to receive a plurality of components, and implementing a carrier aggregation (CA) circuit on the packaging substrate. The CA circuit includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, with the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open-circuit impedance for the second frequency band. The CA circuit further includes a second path implemented between the second filter and the common node, with the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open-circuit impedance for the first frequency band. 
     According to some implementations, the present disclosure relates to a radio-frequency (RF) device having a receiver configured to process RF signals, and an RF module in communication with the receiver. The RF module includes a carrier aggregation (CA) circuit. The CA circuit includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, with the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open-circuit impedance for the second frequency band. The CA circuit further includes a second path implemented between the second filter and the common node, with the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open-circuit impedance for the first frequency band. The RF device further includes an antenna in communication with the RF module, with the antenna being configured to receive the RF signals. 
     In some embodiments, the RF device can be a wireless device. In some embodiments, the wireless device can be a cellular phone. In some embodiments, the antenna can include a diversity antenna, and the RF module can include a diversity receive (DRx) module). The wireless device can further include an antenna switch module (ASM) configured to route the RF signals from the diversity antenna to the receiver. In some embodiments, the DRx module can be implemented between the diversity antenna and the ASM. 
     For purposes of summarizing the disclosure, certain aspects, advantages and novel features of the inventions have been described herein. It is to be understood that not necessarily all such advantages may be achieved in accordance with any particular embodiment of the invention. Thus, the invention may be embodied or carried out in a manner that achieves or optimizes one advantage or group of advantages as taught herein without necessarily achieving other advantages as may be taught or suggested herein. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a carrier aggregation (CA) configuration that includes a CA circuit configured to receive a plurality of inputs and yield an output. 
         FIG. 2  shows a CA aggregation configuration can involve more than two radio-frequency (RF) signals. 
         FIG. 3  shows a more specific example where a CA circuit having one or more features as described herein can be implemented with a low-noise amplifier (LNA) in a receiver. 
         FIG. 4  shows an aggregation configuration where an RF signal is separated at a common input node (RF_IN), and each of the two separated RF signals is processed by a band-pass filter and amplified by an LNA. 
         FIG. 5  shows an example of an aggregation configuration where additional filtering of LNA outputs can be implemented to better isolate the frequency band distributions between different bands. 
         FIG. 6  shows a CA configuration that can be a more specific example of the configuration of  FIG. 3 . 
         FIG. 7  shows the CA configuration of  FIG. 6  being operated in a CA mode. 
         FIG. 8  shows an example CA configuration where the first and second signal paths of  FIG. 7  can be configured to provide selected impedances to facilitate the CA operation. 
         FIG. 9  shows two isolated receive (Rx) paths associated with example bands B3 and B1/4. 
         FIG. 10  shows example Smith plots of complex impedance values for the circuit of  FIG. 9 . 
         FIG. 11  shows two isolated receive (Rx) paths associated with the example bands B3 and B1/4, where each path includes a first phase shifting circuit, a filter, and a second phase shifting circuit between its antenna node and an output node. 
         FIG. 12  shows example Smith plots of complex impedance values for the circuit of  FIG. 11 . 
         FIG. 13  shows the two example receive (Rx) paths of  FIG. 11  connected at the ends so as to yield a common antenna node and a common output node. 
         FIG. 14  shows example Smith plots of complex impedance values for the circuit of  FIG. 13 . 
         FIG. 15  shows various spectrum response curves associated with the examples of  FIGS. 9, 11 and 13 . 
         FIG. 16A  shows examples of the phase shifting circuits of  FIGS. 13-15 . 
         FIG. 16B  shows more examples of the phase shifting circuits of  FIGS. 13-15 . 
         FIG. 17  shows a process that can be implemented to fabricate a device having one or more features as described herein. 
         FIG. 18  shows an RF module having one or more features as described herein. 
         FIG. 19  shows an example of an RF architecture that includes one or more features as described herein. 
         FIG. 20  depicts an example wireless device having one or more advantageous features described herein. 
         FIG. 21  shows that one or more features of the present disclosure can be implemented in a diversity receive module. 
         FIG. 22  shows an example wireless device having the diversity receive module of  FIG. 21 . 
     
    
    
     DETAILED DESCRIPTION OF SOME EMBODIMENTS 
     The headings provided herein, if any, are for convenience only and do not necessarily affect the scope or meaning of the claimed invention. 
     Cellular carrier aggregation (CA) can be supported by allowing two or more radio-frequency (RF) signals to be processed through a common path. For example, carrier aggregation can involve use of a path for a plurality of bands having frequency ranges that are sufficiently separated. In such a configuration, simultaneous operation of more than one band is possible. 
     In the context of a receiver, carrier aggregation can allow concurrent processing of RF signals in a plurality of bands to provide, for example, high data rate capability. In such a carrier aggregation system, it is desirable to maintain a low noise figure (NF) for each RF signal. When two bands being aggregated are close in frequency, maintaining sufficient separation of the two bands is also desirable. 
       FIG. 1  shows a carrier aggregation (CA) configuration  100  that includes a CA circuit  110  configured to receive a plurality of inputs and yield an output. The plurality of inputs can include a first RF signal and a second RF signal. The first RF signal can be provided to the CA circuit  110  from a common input node  102  (RF_IN), through a first path  104   a  that includes a first filter  106   a . Similarly, the second RF signal can be provided to the CA circuit  110  from the common input node  102  (RF_IN), through a second path  104   b  that includes a second filter  106   b . As described herein, the CA circuit  110  can be configured such that the output at a common output node  114  is a recombined RF signal that includes two separated frequency bands associated with the first and second RF signals. As also described herein, the CA circuit  110  can be configured to yield desirable performance features such as low loss, low noise figure, and/or high isolation between the two signal paths  104   a ,  104   b.    
     Various examples herein, including the example of  FIG. 1 , are described in the context of aggregating two frequency bands. However, it will be understood that one or more features of the present disclosure can be implemented in aggregation of more than two frequency bands. For example,  FIG. 2  shows a CA aggregation configuration  100  where three RF signals are separated at a common input node  102  (RF_IN), processed through their respective filters  106   a ,  106   b ,  106   c , and recombined by a CA circuit  110  to yield a recombined RF signal at a common output node  114  (RF_OUT). 
     The aggregation configurations  100  of  FIGS. 1 and 2  can be implemented in a number of RF applications.  FIG. 3  shows a more specific example where a CA circuit  110  having one or more features as described herein can be implemented with a low-noise amplifier (LNA) in a receiver. The CA circuit  110  can be configured to receive a plurality of inputs and yield an output. The plurality of inputs can include a first RF signal and a second RF signal. The first RF signal can be provided to the CA circuit  110  from a common input node  102  (RF_IN), through a first path that includes a first band-pass filter  122 . Similarly, the second RF signal can be provided to the CA circuit  110  from the common input node  102  (RF_IN), through a second path that includes a second band-pass filter  124 . As described herein, the CA circuit  110  can be configured such that the output at a common output node  114  is a recombined RF signal that includes two separated frequency bands associated with the first and second RF signals. As also described herein, the CA circuit  110  can be configured to yield desirable performance features such as low loss, low noise figure, and/or high isolation between the two input signal paths. 
     In  FIG. 3 , the recombined RF signal is shown to be provided to an LNA  130  to amplify and thereby generate a low-noise amplified output signal at an output node  114 . The LNA  130  can be configured to operate with a sufficiently wide bandwidth to effectively amplify the first and second bands of the recombined RF signal. 
     In some embodiments, the pass-band filters  122 ,  124  can be implemented in a number of ways, including, for example, as surface acoustic wave (SAW) filters. It will be understood that other types of filters can be utilized. 
     As described herein, the aggregation configuration  100  of  FIG. 3  can provide a number of advantageous features over other aggregation configurations. For example,  FIG. 4  shows an aggregation configuration  10  where an RF signal is separated at a common input node (RF_IN); and each of the two separated RF signals is processed by a band-pass filter ( 12  or  16 ) and amplified by an LNA ( 14  or  18 ). The separately processed and amplified RF signals (bands “A” and “B”) are shown to be combined by a combiner  20  to yield an output RF signal at a common output node (RF_OUT). 
     In the example of  FIG. 4 , the combined output RF signal includes amplified noise contribution from each of the two LNAs. Accordingly, noise figure can degrade by, for example, approximately 3 dB. 
     Typically, lack of proper isolation between RF paths (e.g., paths associated with bands “A” and “B” of  FIG. 4 ) and their respective bands contributes to the noise figure of the combined RF signal.  FIG. 5  shows an example of an aggregation configuration  30  where additional filtering of LNA outputs can be implemented to better isolate the frequency band distributions between “A” and “B” bands. Similar to the example of  FIG. 4 , the example aggregation configuration  30  includes an RF signal being separated at a common input node (RF_IN); and each of the two separated RF signals being processed by a band-pass filter ( 32  or  42 ) and amplified by an LNA ( 34  or  44 ). The separately processed and amplified RF signals (bands “A” and “B”) are shown to be further filtered by respective filters  36 ,  46  before being combined to yield a combined RF signal at a common output node (RF_OUT). As a result of this filtering, the total noise output at the common output node (RF_OUT) in band “A” typically includes only a contribution from the LNA  34 , while total noise output at the common output node (RF_OUT) in band “B” typically includes only a contribution from the LNA  44 . While this arrangement avoids the aforementioned example 3 dB noise degradation, it typically suffers from the excess cost associated with the two LNAs and the two post-LNA filters. 
     In general, filters constructed from higher Q resonators provide better isolation of frequency bands, especially for bands that are relatively close to each other. For example, cellular frequency bands B1 and B3 have ranges of 2.110 to 2.170 GHz and 1.805 to 1.880 GHz, respectively, for receive operations. For such a pair of relatively close frequency bands, good band isolation is typically not possible with low Q resonators. Accordingly, high Q resonators are typically required or desired. However, use of such additional high Q resonators downstream of the two LNAs (e.g.,  34 ,  44  in  FIG. 5 ) can be undesirable due to, for example, additional cost and required space. 
       FIG. 6  shows a CA configuration  100  that can be a more specific example of the configuration of  FIG. 3 . The CA configuration  100  of  FIG. 6  can provide a number of desirable features, including those that address some or all of the problems associated with the examples of  FIGS. 4 and 5 . 
     In  FIG. 6 , the example CA configuration  100  includes an RF signal being separated at a common input node  102  (RF_IN). The first separated RF signal is shown to be filtered by a band-pass filter  122 , and the second separated RF signal is shown to be filtered by a band-pass filter  124 . The first and second filtered RF signals are shown to be provided to a CA circuit  110  which is configured to yield a combined signal at a common node  126 . 
     The CA circuit  110  is shown to include a phase circuit generally indicated as  150 , and a switch circuit generally indicated as  140 . Examples of functionalities that can be provided by the phase circuit  150  and the switch circuit  140  are described herein in greater detail. 
     The first filtered RF signal from the band-pass filter  122  is shown to be passed through a first phase shifting circuit  152 . Similarly, the second filtered RF signal from the band-pass filter  124  is shown to be passed through a second phase shifting circuit  154 . Examples of such phase shifting circuits are described herein in greater detail. 
     The first and second RF signals from their respective phase shifting circuits ( 152 ,  154 ) are shown to be combined at the common node  126 . In some embodiments, a switch S 1  can be implemented between the first phase shifting circuit  152  and the common node  126 , and a switch S 2  can be implemented between the second phase shifting circuit  154  and the common node  126 . Such switches can allow the CA circuit  110  to operate in a non-CA mode or a CA mode. For example, in  FIG. 6 , the first switch S 1  is shown to be closed, and the second switch S 2  is shown to be open, such that the CA circuit  110  processes the first RF signal in the corresponding frequency band in a non-CA mode. To process the second RF signal in the other frequency band in a non-CA mode, the first switch S 1  can be opened, and the second switch S 2  can be closed. In another example, as shown in  FIG. 7 , both of the first and second switches can be closed, such that the CA circuit  110  processes both of the first and second RF signals in their respective frequency bands in a CA mode. 
     In  FIGS. 6 and 7 , the common node  126  is shown to be coupled to an input of an LNA  120  to allow the processed RF signal (either a combined RF signal in a CA mode or a single-band RF signal in a non-CA mode) to be processed by the LNA  120 . The LNA  120  is shown to generate an amplified RF signal as an output (RF_OUT) at node  114 . 
     In the example of  FIGS. 6 and 7 , the switch circuit  140  can allow the CA circuit  110  to operate in either the non-CA mode or CA mode. In embodiments where the CA circuit  110  is configured to operate in CA mode only, the switch circuit  140  may be omitted. 
       FIG. 8  shows an example CA configuration  100  where the first and second signal paths can be configured to provide selected impedances to facilitate the CA operation. For the purpose of description, such signal paths can be referred to as “A” and “B” bands, and such bands can include any combination of bands suitable for carrier aggregation. As in the example of  FIG. 7 , both of the switches S 1 , S 2  can be in their closed states to facilitate the CA operation. 
     As shown in the example of  FIG. 8 , the first and second phase shifting circuits  152 ,  154  can be utilized to couple the first (A) and second (B) band-pass filters  122 ,  124  with the LNA  120 , and to provide desired impedance values for signals being combined and routed to the LNA  120 . Examples of such adjustments of impedance values by the first and second phase shifting circuits  152 ,  154  are described herein in greater detail. 
     Impedance of the first filter  122  can be tuned to provide a desired impedance for an A-band signal. Accordingly, impedance Z A  for the A-band signal at the output of the A-band filter  122  is approximately at a matched value of Z o  (e.g., 50 Ohms). In the B band, the impedance Z B  for the B-band signal at the output of the A-band filter  122  is not matched to Z o . Since the B band resides in the stopband of the A-band filter, the reflection coefficient |Γ B | of this mismatch is approximately unity. However, the phase of this reflection is typically dependent upon the filter design. Accordingly, the impedance Z B  for the B-band signal at the output of the A-band filter  122  could be any widely mismatched value, either much greater or much smaller than Z o , that results in the condition |Γ B |˜1. 
     Ideally, the A-band filter  122  should present an open circuit for a B-band signal. However, the A-band filter  122  may not provide such an ideal open-circuit impedance for the B-band signal. Accordingly, impedance Z B  for the B-band signal at the output of the A-band filter  122  can be expressed in a complex form Z B =R B +jX B , where the real part (resistance R B ) and the imaginary part (reactance X B ) place the impedance Z B  significantly away from the open circuit state (where one or both of X B  and R B  is/are approximately at infinity). As shown in  FIG. 8 , the first phase shifting circuit  152  can be configured so as to substantially maintain Z o  for Z A , and adjust Z B  from R B  jX B  to, or close to, the open circuit state. 
     Similarly, impedance of the second filter  124  can be tuned to provide a desired impedance for the B-band signal. Accordingly, impedance Z B  for the B-band signal at the output of the B-band filter  124  is approximately at the matched value of Z o  (e.g., 50 Ohms). In the A band, the impedance Z A  for the A-band signal at the output of the B-band filter  122  is not matched to Z o . Since the A band resides in the stopband of the B-band filter, the reflection coefficient |Γ A | of this mismatch is approximately unity. However, the phase of this reflection is dependent upon the filter design. Accordingly, the impedance Z A  for the A-band signal at the output of the B-band filter  122  could be any widely mismatched value, either much greater or much smaller than Z o , that results in the condition |Γ A |˜1. 
     Ideally, the B-band filter  124  should look like an open circuit for the A-band signal. However, the B-band filter  124  may not provide such an ideal open-circuit impedance for the A-band signal. Accordingly, impedance Z A  for the A-band signal at the output of the B-band filter  124  can be expressed in a complex form Z A =R A +jX A , where the real part (resistance R A ) and the imaginary part (reactance X A ) place the impedance Z A  significantly away from the open circuit state (where one or both of X A  and R A  is/are approximately at infinity). As shown in  FIG. 8 , the second phase shifting circuit  154  can be configured so as to substantially maintain Z o  for Z B , and adjust Z A  from R A +jX A  to, or close to, the open circuit state. 
     As shown in  FIG. 8 , the CA configuration  100  with the first and second signal paths configured in the foregoing manner allows the combination of the A-band signal and the B-band signal from their respective paths to be impedance matched for the LNA, and to have the non-band frequency components substantially blocked out. Accordingly, noise figure performance can be improved without having to utilize additional high-Q filters such as SAW filters. 
       FIGS. 9-15  show examples of two frequency band paths and how phase shifting circuits along such paths can yield desirable impedances as described in reference to  FIG. 8 . In  FIG. 9 , two isolated receive (Rx) paths associated with example bands B3 (1,805-1,880 MHz) and B1/4 (2,110-2,170 MHz) are shown. The B3 band path can be considered to be an example of the generic A-band path of  FIG. 8 , and the B1/4 band path can be considered to be an example of the generic B-band path. 
     In  FIG. 9 , the isolated B3 band path includes a B3 filter “A” between an antenna node (ANT B3) and an output node (RX B3). However, the B3 band path does not include a phase shifting circuit between the B3 filter and the output node (RX B3). Similarly, the isolated B1/4 band path includes a B1/4 filter “B” between an antenna node (ANT B1) and an output node (RX B1), but not a phase shifting circuit. 
       FIG. 10  shows example Smith plots of complex impedance values at the RX B1 node (upper left plot), BX B3 node (upper right plot), ANT B1 node (lower left plot), and ANT B3 node (lower right plot) in a frequency sweep between 1.792 GHz and 2.195 GHz, for the circuit of  FIG. 9 . More particularly, impedance values for points m28 and m29 correspond to lower (1.805 GHz) and upper (1.880 GHz) limits, respectively, of the B3 Rx band at the RX B1 node; impedance values for points m6 and m14 correspond to lower (2.110 GHz) and upper (2.170 GHz) limits, respectively, of the B1 Rx band at the RX B3 node; impedance values for points m5 and m7 correspond to lower (1.805 GHz) and upper (1.880 GHz) limits, respectively, of the B3 Rx band at the ANT B1 node; and impedance values for points m3 and m4 correspond to lower (2.110 GHz) and upper (2.170 GHz) limits, respectively, of the B1 Rx band at the ANT B3 node. 
     In  FIG. 10 , one can see that each of the impedance ranges (m28 to m29, m6 to m14, m5 to m7, m3 to m4) is significantly displaced from the open-circuit impedance location on the Smith plot, while remaining close to the outer perimeter of the Smith plot. 
       FIG. 11  shows two isolated receive (Rx) paths associated with the example bands B3 Rx (1,805-1,880 MHz) and B1/4 Rx (2,110-2,170 MHz), where each path includes a first phase shifting circuit, a filter, and a second phase shifting circuit between its antenna node and an output node. More particularly, the B3 path includes a first phase shifting circuit  200 , a B3 filter, and a second phase shifting circuit  202  between an antenna node (ANT B3) and an output node (RX B3). Similarly, the B1/4 band path includes a first phase shifting circuit  204 , a B1/4 filter, and a second phase shifting circuit  206  between an antenna node (ANT B1) and an output node (RX B1). 
       FIG. 12  shows example Smith plots of complex impedance values at the RX B1 node (upper left plot), BX B3 node (upper right plot), ANT B1 node (lower left plot), and ANT B3 node (lower right plot), for the circuit of  FIG. 11 . Similar to the example of  FIG. 10 , impedance values for points m28 and m29 correspond to lower (1.805 GHz) and upper (1.880 GHz) limits, respectively, of the B3 Rx band at the RX B1 node; impedance values for points m6 and m14 correspond to lower (2.110 GHz) and upper (2.170 GHz) limits, respectively, of the B1 Rx band at the RX B3 node; impedance values for points m5 and m7 correspond to lower (1.805 GHz) and upper (1.880 GHz) limits, respectively, of the B3 Rx band at the ANT B1 node; and impedance values for points m3 and m4 correspond to lower (2.110 GHz) and upper (2.170 GHz) limits, respectively, of the B1 Rx band at the ANT B3 node. 
     In  FIG. 12 , one can see that each of the four plots has been rotated, by an amount such that each of the impedance ranges (m28 to m29, m6 to m14, m5 to m7, m3 to m4) straddles the Im(Z)=0 line, and has a large Re(Z) value so as to be at or close to the open-circuit impedance location on the Smith plot. In some embodiments, the amount of delay applied by each delay element in  FIG. 11  can be selected to yield the amount of rotation shown in the corresponding Smith plot. 
     As described in reference to  FIGS. 11 and 12 , it is desirable to have antenna side of the filters to be tuned to present high impedance to opposite-band signals. In some embodiments, the antenna side of a diplexer (e.g., a B3-B1/4 diplexer) can be configured to present a desired high impedance for an opposite-band (e.g., a B3 band frequency signal or a B1/4 band frequency signal) entering a band filter (e.g., a B1/4 band filter or a B3 band filter) of the diplexer. 
       FIG. 13  shows the two example receive (Rx) paths of  FIG. 11  connected at the ends so as to yield a common antenna node (ANT B1/B3) and a common output node (RX B1/B3). In some embodiments, such coupled paths can be configured such that each band path provides a substantially open-circuit impedance for an out-band signal (e.g., B1/4 band signal in the B3 band path, and B3 band signal in the B1/4 band path) as described in  FIGS. 11 and 12 , as well as to provide, when coupled together as shown in  FIG. 13 , matched impedances for the two band signals, at both the common antenna node (ANT B1/B3) and the common output node (RX B1/B3). 
       FIG. 14  shows example Smith plots of complex impedance values at the common output node RX B1/B3 (upper left plot), and at the common antenna node (ANT B1/B3) (lower left plot), for the circuit of  FIG. 13 . For the RX B1/B3 node, impedance values for points m53 and m54 correspond to B3 band frequencies of 1.805 GHz and 1.880 GHz, respectively, and impedance values for points m55 and m56 correspond to B1 band frequencies of 2.110 GHz and 2.170 GHz, respectively. All points m54, m54, m55, m56 are clustered near the center of the Smith chart, indicating that the impedance of the RX B1/B3 node is substantially well-matched to 50 ohms at all frequencies in both bands B1 and B3. This occurs because each path in its own band is generally undisturbed by the other path; thus the combined circuit presents a match in band B1 determined substantially by the B1 path alone, and a match in band B3 determined substantially by the B3 path alone, even though the paths are physically tied together. 
     Likewise, for the ANT B1/B3 node, impedance values for points m46 and m47 correspond to frequencies of 1.805 GHz and 1.880 GHz, respectively, and impedance values for points m48 and m49 correspond to B1 band frequencies of 2.110 GHz and 2.170 GHz, respectively. All points m46, m47, m48, m49 are clustered near the center of the Smith chart, indicating that the impedance of the ANT B1/B3 node is substantially well-matched to 50 ohms at all frequencies in both bands B1 and B3. This occurs because each path in its own band is generally undisturbed by the other path; thus the combined circuit presents a match in band B1 determined substantially by the B1 path alone, and a match in band B3 determined substantially by the B3 path alone, even though the paths are physically tied together. 
       FIG. 14  further shows a distribution of reflection coefficient S 11  (S( 4 , 4 ) in  FIG. 14 ) at the ANT B1/B3 node (lower right panel) and a distribution of reflection coefficient S 22  (S( 5 , 5 ) in  FIG. 14 ) at the RX B1/B3 node (upper right panel), for the circuit of  FIG. 13 . In the S 11  (S( 4 , 4 )) and S 22  (S( 5 , 5 )) distributions, the matching at each of the two RX bands (B3 RX and B1 RX) is prominent 
       FIG. 15  shows, in the upper left panel, a spectrum response of the B3 receive path, and an independent spectrum response of the B1 receive path, for the circuit of  FIG. 9 . The same responses apply, substantially unchanged, to the circuit of  FIG. 11 , since the added delays in  FIG. 11  generally affect only phase and not amplitude for each path. A peak in gain of the B3 RX band (e.g., a B3 passband) is indicated as  230 , and a peak in gain of the B1 RX band (e.g., a B1 passband) is indicated as  232 . It is noted that each path exhibits greater than 30 dB attenuation in the opposite band. 
     In  FIG. 15 , the upper right panel shows a single spectrum response for the circuit of  FIG. 13 . It is noted that this single response exhibits two passbands, with the B3 RX passband indicated as  234 , and the B1 RX passband indicated as  236 . 
     In  FIG. 15 , the lower left panel shows an overlap of the B3 RX passband  230  exhibited by the independent B3 receive path of  FIG. 9 / FIG. 11 , with the B3 RX passband  234  exhibited by the combined circuit of  FIG. 13 . The lower right panel shows an overlap of the B1 RX passband  232  exhibited by the independent B1 receive path of  FIG. 9 / FIG. 11 , with the B1 RX passband  236  exhibited by the combined circuit of  FIG. 13 . In both examples of the lower right panel and the lower left panel, one can see that each passband of the combined circuit of  FIG. 13  substantially resembles the passband of the respective independent receive path from which it was formed, in both bandwidth and characteristic ripple. Further, the gain distribution of a given band before the corresponding phase shifting circuit is added and the paths combined, is only slightly higher than the distribution after such addition of the phase shifting circuit and combining of the paths. Thus, one can see that the phase shifting circuits having one or more features as described herein can be configured to provide desired functionalities with little or no loss. 
       FIG. 16A  shows more specific examples of the phase shifting circuits such as circuits  212 ,  216  of  FIGS. 13-15 . In  FIG. 16A , a CA configuration  100  can be an example of the CA configuration  100  of  FIG. 8 . An input RF signal (RF_IN) from an antenna can be received at an input node  102 . A diplexer  260  is shown to be coupled to the input node  102  so as to receive the input RF signal. Such a received signal can be processed through filters  122 ,  124  configured to provide band-pass functionality for band A and band B. Examples of such bands are described herein in greater detail. In some embodiments, the diplexer  260  can be configured to provide impedance matching for the input RF signal, as described herein in reference to  FIGS. 11-15 . 
     Outputs of the band-pass filters  122 ,  124  are shown to be routed to a first path that includes a first phase shifting circuit  152 , and a second path that includes a second phase shifting circuit  154 . The first path is shown to further include a switch S 1  between the first phase shifting circuit  152  and a common output node. The second path is shown to further include a switch S 2  between the second phase shifting circuit  154  and the common output node. 
     The common output node receiving processed signals from the foregoing first and second paths is shown to be coupled to an input of an LNA  120 . The LNA  120  is shown to yield an amplified output signal (RF_OUT) at a node  114 . 
     The first phase shifting circuit  152  is shown to include capacitances C 5  and C 6  arranged in series between its input (from an output of the band A filter  122 ) and the switch S 1 . An inductance L 5  is shown to couple a node between C 5  and C 6  with ground. 
     The second phase shifting circuit  154  is shown to include capacitances C 2  and C 3  arranged in series between its input (from an output of the band B filter  124 ) and the switch S 2 . An inductance L 4  is shown to couple a node between C 2  and C 3  with ground. 
       FIG. 16B  shows more specific examples of the phase shifting circuits such as circuits  212 ,  216  of  FIGS. 13-15 , implemented in a low-pass phase shifter configuration. In  FIG. 16B , a CA configuration  100  can be an example of the CA configuration  100  of  FIG. 8 . An input RF signal (RF_IN) from an antenna can be received at an input node  102 . A diplexer  260  is shown to be coupled to the input node  102  so as to receive the input RF signal. Such a received signal can be processed through filters  122 ,  124  configured to provide band-pass functionality for band A and band B. Examples of such bands are described herein in greater detail. In some embodiments, the diplexer  260  can be configured to provide impedance matching for the input RF signal, as described herein in reference to  FIGS. 11-15 . 
     Outputs of the band-pass filters  122 ,  124  are shown to be routed to a first path that includes a first phase shifting circuit  152 , and a second path that includes a second phase shifting circuit  154 . The first path is shown to further include a switch S 1  between the first phase shifting circuit  152  and a common output node. The second path is shown to further include a switch S 2  between the second phase shifting circuit  154  and the common output node. 
     The common output node receiving processed signals from the foregoing first and second paths is shown to be coupled to an input of an LNA  120 . The LNA  120  is shown to yield an amplified output signal (RF_OUT) at a node  114 . 
     The first phase shifting circuit  152  is shown to include inductances L 5 ′ and L 6 ′ arranged in series between its input (from an output of the band A filter  122 ) and the switch S 1 . A capacitance C 5 ′ is shown to couple a node between L 5 ′ and L 6 ′ with ground. 
     The second phase shifting circuit  154  is shown to include inductances L 2 ′ and L 3 ′ arranged in series between its input (from an output of the band B filter  124 ) and the switch S 2 . A capacitance C 4 ′ is shown to couple a node between L 2 ′ and L 3 ′ with ground. 
     In some embodiments, various functionalities as described herein in reference to, for example,  FIGS. 8 and 13  can be obtained with values of capacitances and inductances for the example configuration of  FIG. 16A  for the example bands of B3 RX and B1/4 RX, as listed in Table 1. 
                                 TABLE 1                       Capacitance/inductance   Approximate value                                                        C2   1.71   pF           C3   1.71   pF           C5   5.38   pF           C6   6.38   pF           L4   4   nH           L5   7.668   nH                        
It will be understood that for other pairs of bands, values for the capacitances and inductances can be selected accordingly. It will also be understood that the same or similar various functionalities may be accomplished with appropriate values for the elements of the example circuit of  FIG. 16B .
 
     In some embodiments, some or all of the capacitances and/or inductances can be implemented as parts of signal paths or other conductive features, as lumped elements, or any combination thereof. 
       FIG. 17  shows a process  280  that can be implemented to fabricate a device having one or more features as described herein. In block  282 , a circuit having at least a diplexer functionality can be mounted or provided on a substrate. In various examples, carrier aggregation (CA) is described in the context of diplexers; however, it will be understood that CA can also be implemented with more than two bands (e.g., utilizing multiplexers). In some embodiments, a diplexer can be implemented as a device; and such a device can be mounted on the substrate. 
     In block  284 , a first phase shifting circuit can be formed or provided between a first output of the diplexer circuit and an input of a first switch. In block  286 , a second phase shifting circuit can be formed or provided between a second output of the diplexer circuit and an input of a second switch. In block  288 , an output of the first switch and an output of the second switch can be coupled with a common node. In some embodiments, such a configuration of the first and second phase shifting circuits being coupled to the common node through their respective switches can facilitate operation of the device in a CA mode or a non-CA mode. 
     In block  290 , the common node can be coupled to an input of a low-noise amplifier (LNA). In some embodiments, such an aggregation of the two signal paths into a single LNA can allow the LNA to operate in the CA mode or the non-CA mode, as determined by the state of the switches. 
     In some embodiments, the device described in  FIG. 17  can be a module configured for RF applications.  FIG. 18  shows a block diagram of an RF module  300  (e.g., a front-end module) having a packaging substrate  302  such as a laminate substrate. Such a module can include one or more LNAs; and in some embodiments, such LNA(s) can be implemented on a semiconductor die  304 . An LNA implemented on such a die can be configured to receive RF signals through signal paths as described herein. Such an LNA can also benefit from the one or more advantageous features associated with improved carrier aggregation (CA) functionalities as described herein. 
     The module  300  can further include a plurality of switches implemented on one or more semiconductor die  306 . Such switches can be configured to provide the various switching functionalities as described herein, including providing and/or facilitating isolation, enabling/disabling CA mode of operation, and band selection in a non-CA mode. 
     The module  300  can further include one or more diplexers and/or a plurality of filters (collectively indicated as  310 ) configured to process RF signals. Such diplexers/filters can be implemented as surface-mount devices (SMDs), as part of an integrated circuit (IC), of some combination thereof. Such diplexers/filters can include or be based on, for example, SAW filters, and can be configured as high Q devices. 
     In  FIG. 18 , a plurality of phase shifting circuits are collectively indicated as  308 . Such phase shifting circuits can include one or more features as described herein to provide, among others, improved isolation between paths associated with different bands being operated in a CA mode. 
       FIG. 19  shows an example of an RF architecture  400  that includes one or more features as described herein. In some embodiments, such an architecture can be implemented on a module  300  such as the example described in reference to  FIG. 18 . It will be understood that the architecture  400  of  FIG. 19  does not necessarily need to be confined to module. 
     The example architecture  400  of  FIG. 19  can include a number of signal paths configured for receiving and/or transmitting RF signals. The architecture  400  can also include an antenna switching circuit  404  coupled to an antenna port  402 . Such an antenna switching circuit can be configured to route RF signals in cellular frequency ranges to multiple paths associated with different cellular bands. In the example shown, the antenna switching circuit  404  includes a single-pole-2-throw (SP2T) switch, with the pole being coupled to the antenna port  402 . 
     In the context of the example RX paths, the first path is configured for B2/625/4 bands, and the second path is configured for B3/61/4 bands. RF signals associated with such bands are shown to be processed by their respective filters  406 . 
     Signals in the B2/625/4 bands (e.g., 1.930 to 1.995 GHz and 2.110 to 2.155 GHz) of the first path can be carrier aggregated as described herein and be amplified by an LNA of the group of LNAs  410 . As described herein, carrier aggregation for the B2/625/4 bands can include a plurality of phase shifting circuits implemented between the B2/625/4 diplexer and the LNA. As also described herein, the paths between such phase shifting circuits and the LNA can include respective switches to allow operations in CA mode as well as non-CA mode. 
     Signals in the B3/B1/4 bands (e.g., 1.805 to 1.880 GHz and 2.110 to 2.170 GHz) of the second path can be carrier aggregated as described herein and be amplified by an LNA of the group of LNAs  410 . Such an LNA can be configured to provide bandwidth coverage of, for example, 1.805 to 2.170 GHz. As described herein, such carrier aggregation can include a plurality of phase shifting circuits implemented between the B3/B1/4 diplexer and the LNA. As also described herein, the paths between such phase shifting circuits and the LNA can include respective switches to allow operations in CA mode as well as non-CA mode. 
     The amplified signals from the LNA can be routed to a band selection switch  412 . The band selection switch  412  is shown to be coupled to a node  416  to allow further processing of an amplified RF signal from the selected LNA. 
     In some implementations, an architecture, device and/or circuit having one or more features described herein can be included in an RF device such as a wireless device. Such an architecture, device and/or circuit can be implemented directly in the wireless device, in one or more modular forms as described herein, or in some combination thereof. In some embodiments, such a wireless device can include, for example, a cellular phone, a smart-phone, a hand-held wireless device with or without phone functionality, a wireless tablet, a wireless router, a wireless access point, a wireless base station, etc. Although described in the context of wireless devices, it will be understood that one or more features of the present disclosure can also be implemented in other RF systems such as base stations. 
       FIG. 20  schematically depicts an example wireless device  500  having one or more advantageous features described herein. In some embodiments, such advantageous features can be implemented in a front-end (FE) module  300  and/or in an architecture  400  as described herein. One or more of such features can also be implemented in a main antenna switch module (ASM)  514 . In some embodiments, such an FEM/architecture can include more or less components than as indicated by the dashed box. 
     PAs in a PA module  512  can receive their respective RF signals from a transceiver  510  that can be configured and operated to generate RF signals to be amplified and transmitted, and to process received signals. The transceiver  510  is shown to interact with a baseband sub-system  508  that is configured to provide conversion between data and/or voice signals suitable for a user and RF signals suitable for the transceiver  510 . The transceiver  510  is also shown to be connected to a power management component  506  that is configured to manage power for the operation of the wireless device  500 . Such power management can also control operations of the baseband sub-system  508  and other components of the wireless device  500 . 
     The baseband sub-system  508  is shown to be connected to a user interface  502  to facilitate various input and output of voice and/or data provided to and received from the user. The baseband sub-system  508  can also be connected to a memory  504  that is configured to store data and/or instructions to facilitate the operation of the wireless device, and/or to provide storage of information for the user. 
     In the example wireless device  500 , the front-end module  300 /architecture  400  can include one or more carrier aggregation-capable signal paths configured to provide one or more functionalities as described herein. Such signal paths can be in communication with an antenna switch module (ASM)  404  through their respective diplexer(s). In some embodiments, at least some of the signals received through a diversity antenna  530  can be routed from the ASM  404  to one or more low-noise amplifiers (LNAs)  518  in manners as described herein. Amplified signals from the LNAs  518  are shown to be routed to the transceiver  510 . 
     A number of other wireless device configurations can utilize one or more features described herein. For example, a wireless device does not need to be a multi-band device. In another example, a wireless device can include additional antennas such as diversity antenna, and additional connectivity features such as Wi-Fi, Bluetooth, and GPS. 
     Examples Related to Diversity Receive (DRx) Implementation: 
     Using one or more main antennas and one or more diversity antennas in a wireless device can improve quality of signal reception. For example, a diversity antenna can provide additional sampling of RF signals in the vicinity of the wireless device. Additionally, a wireless device&#39;s transceiver can be configured to process the signals received by the main and diversity antennas to obtain a receive signal of higher energy and/or improved fidelity, when compared to a configuration using only the main antenna. 
     To reduce the correlation between signals received by the main and diversity antennas and/or to enhance antenna isolation, the main and diversity antennas can be separated by a relatively large physical distance in the wireless device. For example, the diversity antenna can be positioned near the top of the wireless device and the main antenna can be positioned near the bottom of the wireless device, or vice-versa. 
     The wireless device can transmit or receive signals using the main antenna by routing corresponding signals from or to the transceiver through an antenna switch module. To meet or exceed design specifications, the transceiver, the antenna switch module, and/or the main antenna can be in relatively close physical proximity to one another in the wireless device. Configuring the wireless device in this manner can provide relatively small signal loss, low noise, and/or high isolation. 
     In the foregoing example, the main antenna being physically close to the antenna switch module can result in the diversity antenna being positioned relatively far from the antenna switch module. In such a configuration, a relatively long signal path between the diversity antenna and the antenna switch module can result in significant loss and/or addition of loss associated with the signal received through the diversity antenna. Accordingly, processing of the signal received through the diversity antenna, including implementation of one or more features as described herein, in the close proximity to the diversity antenna can be advantageous. 
       FIG. 21  shows that in some embodiments, one or more features of the present disclosure can be implemented in a diversity receive (DRx) module  300 . Such a module can include a packaging substrate  302  (e.g., a laminate substrate) configured to receive a plurality of components, as well to provide or facilitate electrical connections associated with such components. 
     In the example of  FIG. 21 , the DRx module  300  can be configured to receive an RF signal from a diversity antenna (not shown in  FIG. 21 ) at an input  320  and route such an RF signal to a low-noise amplifier (LNA)  332 . It will be understood that such routing of the RF signal can involve carrier-aggregation (CA) and/or non-CA configurations. It will also be understood that although one LNA (e.g., a broadband LNA) is shown, there may be more than one LNAs in the DRx module  300 . Depending on the type of LNA and the mode of operation (e.g., CA or non-CA), an output  334  of the LNA  332  can include one or more frequency components associated with one or more frequency bands. 
     In some embodiments, some or all of the foregoing routing of the RF signal between the input  320  and the LNA  332  can be facilitated by an assembly of one or more switches  322  between the input  320  and an assembly of diplexer(s) and/or filter(s) (collectively indicated as  324 ), and an assembly of one or more switches  330  between the diplexer/filter assembly  324  and the LNA  332 . In some embodiments, the switch assemblies  322 ,  330  can be implemented on, for example, one or more silicon-on-insulator (SOI) die. In some embodiments, some or all of the foregoing routing of the RF signal between the input  320  and the LNA  332  can be achieved without some or all of the switches associated with the switch-assemblies  322 ,  330 . 
     In the example of  FIG. 21 , the diplexer/filter assembly  324  is depicted as including two example diplexers  326  and two individual filters  328 . It will be understood that the DRx module  300  can have more or less numbers of diplexers, and more or less numbers of individual filters. Such diplexer(s)/filter(s) can be implemented as, for example, surface-mount devices (SMDs), as part of an integrated circuit (IC), of some combination thereof. Such diplexers/filters can include or be based on, for example, SAW filters, and can be configured as high Q devices. 
     In some embodiments, the DRx module  300  can include a control component such as a MIPI RFFE interface  340  configured to provide and/or facilitate control functionalities associated with some or all of the switch assemblies  322 ,  330  and the LNA  332 . Such a control interface can be configured to operate with one or more I/O signals  342 . 
       FIG. 22  shows that in some embodiments, a DRx module  300  having one or more features as described herein (e.g., DRx module  300  of  FIG. 21 ) can be included in an RF device such as a wireless device  500 . In such a wireless device, components such as user interface  502 , memory  504 , power management  506 , baseband sub-system  508 , transceiver  510 , power amplifier (PA)  512 , antenna switch module (ASM)  514 , and antenna  520  can be generally similar to the examples of  FIG. 20 . 
     In some embodiments, the DRx module  300  can be implemented between one or more diversity antennas and the ASM  514 . Such a configuration can allow an RF signal received through the diversity antenna  530  to be processed (in some embodiments, including amplification by an LNA) with little or no loss of and/or little or no addition of noise to the RF signal from the diversity antenna  530 . Such processed signal from the DRx module  300  can then be routed to the ASM through one or more signal paths  532  which can be relatively lossy. 
     In the example of  FIG. 22 , the RF signal from the DRx module  300  can be routed through the ASM  514  to the transceiver  510  through one or more receive (Rx) paths. Some or all of such Rx paths can include their respective LNA(s). In some embodiments, the RF signal from the DRx module  300  may or may not be further amplified with such LNA(s). 
     One or more features of the present disclosure can be implemented with various cellular frequency bands as described herein. Examples of such bands are listed in Table 2. It will be understood that at least some of the bands can be divided into sub-bands. It will also be understood that one or more features of the present disclosure can be implemented with frequency ranges that do not have designations such as the examples of Table 2. 
     
       
         
           
               
               
               
               
               
             
               
                   
                 TABLE 2 
               
               
                   
                   
               
               
                   
                   
                   
                 Tx Frequency 
                 Rx Frequency 
               
               
                   
                 Band 
                 Mode 
                 Range (MHz) 
                 Range (MHz) 
               
               
                   
                   
               
             
            
               
                   
                 B1 
                 FDD 
                 1,920-1,980 
                 2,110-2,170 
               
               
                   
                 B2 
                 FDD 
                 1,850-1,910 
                 1,930-1,990 
               
               
                   
                 B3 
                 FDD 
                 1,710-1,785 
                 1,805-1,880 
               
               
                   
                 B4 
                 FDD 
                 1,710-1,755 
                 2,110-2,155 
               
               
                   
                 B5 
                 FDD 
                 824-849 
                 869-894 
               
               
                   
                 B6 
                 FDD 
                 830-840 
                 875-885 
               
               
                   
                 B7 
                 FDD 
                 2,500-2,570 
                 2,620-2,690 
               
               
                   
                 B8 
                 FDD 
                 880-915 
                 925-960 
               
               
                   
                 B9 
                 FDD 
                 1,749.9-1,784.9 
                 1,844.9-1,879.9 
               
               
                   
                 B10 
                 FDD 
                 1,710-1,770 
                 2,110-2,170 
               
               
                   
                 B11 
                 FDD 
                 1,427.9-1,447.9 
                 1,475.9-1,495.9 
               
               
                   
                 B12 
                 FDD 
                 699-716 
                 729-746 
               
               
                   
                 B13 
                 FDD 
                 777-787 
                 746-756 
               
               
                   
                 B14 
                 FDD 
                 788-798 
                 758-768 
               
               
                   
                 B15 
                 FDD 
                 1,900-1,920 
                 2,600-2,620 
               
               
                   
                 B16 
                 FDD 
                 2,010-2,025 
                 2,585-2,600 
               
               
                   
                 B17 
                 FDD 
                 704-716 
                 734-746 
               
               
                   
                 B18 
                 FDD 
                 815-830 
                 860-875 
               
               
                   
                 B19 
                 FDD 
                 830-845 
                 875-890 
               
               
                   
                 B20 
                 FDD 
                 832-862 
                 791-821 
               
               
                   
                 B21 
                 FDD 
                 1,447.9-1,462.9 
                 1,495.9-1,510.9 
               
               
                   
                 B22 
                 FDD 
                 3,410-3,490 
                 3,510-3,590 
               
               
                   
                 B23 
                 FDD 
                 2,000-2,020 
                 2,180-2,200 
               
               
                   
                 B24 
                 FDD 
                 1,626.5-1,660.5 
                 1,525-1,559 
               
               
                   
                 B25 
                 FDD 
                 1,850-1,915 
                 1,930-1,995 
               
               
                   
                 B26 
                 FDD 
                 814-849 
                 859-894 
               
               
                   
                 B27 
                 FDD 
                 807-824 
                 852-869 
               
               
                   
                 B28 
                 FDD 
                 703-748 
                 758-803 
               
               
                   
                 B29 
                 FDD 
                 N/A 
                 716-728 
               
               
                   
                 B30 
                 FDD 
                 2,305-2,315 
                 2,350-2,360 
               
               
                   
                 B31 
                 FDD 
                 452.5-457.5 
                 462.5-467.5 
               
               
                   
                 B33 
                 TDD 
                 1,900-1,920 
                 1,900-1,920 
               
               
                   
                 B34 
                 TDD 
                 2,010-2,025 
                 2,010-2,025 
               
               
                   
                 B35 
                 TDD 
                 1,850-1,910 
                 1,850-1,910 
               
               
                   
                 B36 
                 TDD 
                 1,930-1,990 
                 1,930-1,990 
               
               
                   
                 B37 
                 TDD 
                 1,910-1,930 
                 1,910-1,930 
               
               
                   
                 B38 
                 TDD 
                 2,570-2,620 
                 2,570-2,620 
               
               
                   
                 B39 
                 TDD 
                 1,880-1,920 
                 1,880-1,920 
               
               
                   
                 B40 
                 TDD 
                 2,300-2,400 
                 2,300-2,400 
               
               
                   
                 B41 
                 TDD 
                 2,496-2,690 
                 2,496-2,690 
               
               
                   
                 B42 
                 TDD 
                 3,400-3,600 
                 3,400-3,600 
               
               
                   
                 B43 
                 TDD 
                 3,600-3,800 
                 3,600-3,800 
               
               
                   
                 B44 
                 TDD 
                 703-803 
                 703-803 
               
               
                   
                   
               
            
           
         
       
     
     For the purpose of description, it will be understood that “multiplexer,” “multiplexing” and the like can include “diplexer,” “diplexing” and the like. 
     Unless the context clearly requires otherwise, throughout the description and the claims, the words “comprise,” “comprising,” and the like are to be construed in an inclusive sense, as opposed to an exclusive or exhaustive sense; that is to say, in the sense of “including, but not limited to.” The word “coupled”, as generally used herein, refers to two or more elements that may be either directly connected, or connected by way of one or more intermediate elements. Additionally, the words “herein,” “above,” “below,” and words of similar import, when used in this application, shall refer to this application as a whole and not to any particular portions of this application. Where the context permits, words in the above Detailed Description using the singular or plural number may also include the plural or singular number respectively. The word “or” in reference to a list of two or more items, that word covers all of the following interpretations of the word: any of the items in the list, all of the items in the list, and any combination of the items in the list. 
     The above detailed description of embodiments of the invention is not intended to be exhaustive or to limit the invention to the precise form disclosed above. While specific embodiments of, and examples for, the invention are described above for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the relevant art will recognize. For example, while processes or blocks are presented in a given order, alternative embodiments may perform routines having steps, or employ systems having blocks, in a different order, and some processes or blocks may be deleted, moved, added, subdivided, combined, and/or modified. Each of these processes or blocks may be implemented in a variety of different ways. Also, while processes or blocks are at times shown as being performed in series, these processes or blocks may instead be performed in parallel, or may be performed at different times. 
     The teachings of the invention provided herein can be applied to other systems, not necessarily the system described above. The elements and acts of the various embodiments described above can be combined to provide further embodiments. 
     While some embodiments of the inventions have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the disclosure. Indeed, the novel methods and systems described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the methods and systems described herein may be made without departing from the spirit of the disclosure. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the disclosure.