Patent Publication Number: US-9891762-B2

Title: Touch screen device and method for driving the same

Description:
This application claims the priority benefit of Korean Patent Application No. 10-2013-0104344 filed on Aug. 30, 2013, which is incorporated herein by reference for all purposes as if fully set forth herein. 
     BACKGROUND 
     Field 
     This document relates to a touch screen device and a method for driving the same. 
     Related Art 
     A user interface (UI) enables the communication of a user with various kinds of electric and electronic devices, so that the user can easily control the devices. Representative examples of the user interface are a keypad, a keyboard, a mouse, an on-screen display (OSD), a remote controller having an infrared communication or radio frequency (RF) communication function, and the like. The user interface technology has continuously developed to improve the user&#39;s sensitivity and ease of operation. The user interface has evolved into a touch UI, a voice recognition UI, a 3D UI, etc. The touch UI tends to be adopted in portable information devices, but has also expanded to electronic home appliances. 
     As one example of a touch screen for implementing the touch UI, a mutual capacitance type touch screen device that can respectively recognize multiple touches is gaining popularity. 
     The mutual capacitance type touch screen device includes a touch screen panel having Tx lines, Rx lines crossing the Tx lines, and touch sensors formed at crossings of the Tx lines and the Rx lines. Each of the touch sensors has mutual capacitance. The touch screen device senses a charge variation of each of the touch sensors between the time before and after a touch to determine the touch or non-touch with a conductive material and the position of touch. The touch screen device calculates touch coordinates by supplying a driving pulse to the Tx lines of the touch screen panel, converting charge variations of the touch sensors, which are received through the Rx lines, into touch raw data which are digital data, and analyzing the touch raw data. 
     The touch screen device can minimize an erroneous calculation by removing noise of the charge variations of the touch sensors. However, the range of the noise of the charge variations of the touch sensors may be slightly different according to different products. Due to this, a different noise filter for removing the noise of the charge variations of the touch sensors needs to be used depending on specific products. 
     SUMMARY 
     The present invention has been made in an effort to provide a touch screen device including a noise filter capable of changing a pass band, and a method for driving the same. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings: 
         FIG. 1  is a block diagram schematically showing a display device and a touch screen device according to a first embodiment; 
         FIG. 2  is a flowchart showing a method for driving the touch screen device according to the first embodiment; 
         FIG. 3  is a block diagram specifically showing an Rx driving circuit in  FIG. 1 ; 
         FIG. 4  is a flowchart showing a method for driving an Rx driving circuit according to an embodiment; 
         FIG. 5  is a waveform diagram showing outputs of a noise filter, an integrator, and a sampling circuit; 
         FIG. 6  is a graph showing pass bands of a noise filter; 
         FIG. 7  is a circuit diagram specifically showing a noise filter according to a first embodiment; 
         FIGS. 8A and 8B  illustrate graphs showing pass band widths of a noise filter for different Q-factors; 
         FIG. 9A  shows exemplary resistance values of variable resistors and capacitance values of variable capacitors when the Q-factor is 0.5; 
         FIG. 9B  shows exemplary resistance values of variable resistors and capacitance values of variable capacitors when the Q-factor is 2; 
         FIGS. 10A and 10B  are exemplary views showing variable resistors and variable capacitors; and 
         FIG. 11  is a circuit diagram showing a noise filter according to a second embodiment. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     Reference will now be made in detail to embodiments of the invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers will be used throughout the drawings to refer to the same or like parts. It will be noted that detailed description of known art will be omitted if it is determined that the art can confuse an understanding of the embodiments of the invention. 
       FIG. 1  is a block diagram schematically showing a display device and a touch screen device according to a first embodiment. Referring to  FIG. 1 , a display device includes a display panel DIS, a gate driving circuit  10 , a data driving circuit  20 , a timing controller  30 , a host system  70 , etc. A touch screen device includes a touch screen panel (TSP), a touch driving circuit  40 , a touch coordinate calculation unit  50 , etc. 
     The display device according to an embodiment may be implemented as a flat panel display device, such as liquid crystal display (LCD), field emission display (FED), plasma display panel (PDP), organic light emitting display (OLED), or electrophoresis (EPD). In the following description, the present disclosure will be described based on a liquid crystal display that implements a display device according to an embodiment, but it is noted that the present invention is not limited thereto. 
     The display panel DIS includes a liquid crystal layer formed between a lower substrate and an upper substrate. A plurality of data lines D 1  to Dm (m is a natural number) and a plurality of gate lines G 1  to Gn (n is a natural number) crossing the data lines D 1  to Dm are formed on the lower substrate of the display panel DIS. Also formed are: a plurality of thin film transistors at crossings of the data lines D 1  to Dm and the gate lines G 1  to Gn, a plurality of pixel electrodes for charging liquid crystal cells to data voltages, a plurality of storage capacitors connected to the plurality of pixel electrodes to hold a voltage of the liquid crystal cells, etc. 
     A black matrix, color filters, etc., may be formed on the upper substrate of the display panel DIS. However, in the case where the display panel DIS has a color filter on the TFT (COT) structure, the black matrix and the color filter may be formed on the lower substrate of the display panel DIS. The display panel DIS may be implemented in any known mode including a twisted nematic (TN) mode, a vertical alignment (VA) mode, an in-plane switching (IPS) mode, and a fringe field switching (FFS) mode. 
     Polarizing plates are respectively attached to the upper and lower substrates GLS 1  and GLS 2  of the display panel DIS. Alignment layers for setting a pre-tilt angle of liquid crystals are respectively formed on inner surfaces of the upper and lower substrates, which are contacted with liquid crystals. Column spacers for maintaining cell gaps of the liquid crystal cells are formed between the upper and lower substrates the display panel DIS. A backlight unit may be disposed below a rear surface of the display panel DIS. The backlight unit may be implemented in an edge type backlight unit and a direct type backlight unit to provide light to the display panel DIS. 
     The data driving circuit  20  receives digital image data RGB and a source timing control signal DSC from the timing controller  30 . The data driving circuit  20  converts the digital video data RGB into positive/negative data voltages in response to the source timing controller DSC, and supplies the data voltages to the data lines. The gate driving circuit  10  sequentially supplies gate pulses (or scan pulses) synchronized with the data voltages to the gate lines G 1  to Gn to select pixels of the display panel DIS to which the data voltages are supplied. 
     The timing controller  30  receives the digital image data RGB and timing signals from the host system  70 . The timing signals may include a vertical sync signal, a horizontal sync signal, a data enable signal, a dot clock, etc. The vertical synchronization signal is a signal that defines one frame period. The horizontal synchronization signal is a signal that defines one horizontal period necessary to supply the data voltages to the pixels in one horizontal line in the display panel DIS. The data enable signal is a signal that defines a period during which effective data are input. The dot clock is a signal that repeats with a short cycle time. 
     In order to control the operation timings of the gate driving circuit  10  and the data driving circuit  20 , the timing controller  30  generates a source timing control signal DCS for controlling the operation timing of the data driving circuit  20  and a gate timing signal GCS for controlling the operation timing of the gate driving circuit  10  based on timing signals. The timing controller  30  outputs the gate timing control signal GCS to the gate driving circuit  10 , and outputs the digital image data RGB and the source timing control signal DCS to the data driving circuit  20 . 
     The host system  70  may be implemented as any one of a navigation system, a set-top box, a DVD player, a Blu-ray disk player, a personal computer (PC), a home theater system, a broadcast receiver, and a phone system. The host system  70  includes a system on chip (SoC) with a built-in scaler to convert the digital image data RGB of an input image into a format suitable for display on the display panel DIS. The host system  70  transmits the digital image data RGB and the timing signals to the timing controller  30 . 
     The touch screen device according to an embodiment will now be described in detail. A touch screen panel TSP includes Tx lines T 1  to Tj (j is a natural number of 2 or greater), Rx lines R 1  to Ri (i is a natural number of 2 or greater) crossing the Tx lines T 1  to Tj, and i×j touch sensors formed at crossings of the Tx lines T 1  to Tj and Rx lines R 1  to Ri. The respective touch sensors may be implemented to have mutual capacitance in terms of an equivalent circuit, but it is noted that the touch sensors are not limited thereto. 
     In the case where the touch screen device is combined with the display device, the touch screen panel TSP may be joined to an upper portion of the display panel DIS. Particularly, in the case where the display device is implemented as the liquid crystal display, the touch screen panel TSP may be joined onto an upper polarizing plate of the display panel DIS or joined between the upper polarizing plate and the upper substrate of the display panel DIS. In addition, the touch sensors of the touch screen panel TSP may be formed on the lower substrate within the display panel together with a pixel array (in an in-cell type). 
     The touch driving circuit  40  supplies the driving pulse to the Tx lines T 1  to Tj, and is synchronized by the driving pulse to sense the charge variations of the respective touch sensors through the Rx lines R 1  to Ri. The touch driving circuit  40  includes a Tx driving circuit  41 , an Rx driving circuit  42 , and a touch controller  43 . The Tx driving circuit  41 , the Rx driving circuit  42 , and the touch controller  43  may be integrated in one read-out IC (ROIC). 
       FIG. 2  is a flowchart showing a method for driving the touch screen device according to the first embodiment. The method for driving the touch screen device according to the first embodiment will be described in detail below. 
     The Tx driving circuit  41  selects a Tx line, to which the driving pulse is to be output, under the control of the touch controller  43 , and supplies the driving pulse to the selected Tx line (S 101 ). The Rx driving circuit  42  selects Rx lines, which are to receive charge variations of the touch sensors, under the control of the touch controller  43 , and receives the charge variations of the touch sensors through the selected Rx lines (S 102 ). The Rx driving circuit  42  samples the charge variations of the touch sensors, which are received through the Rx lines R 1  to Ri, and converts the received charge variations into touch raw data TRD (S 103 ). The Rx driving circuit  42  and a driving method thereof will be described later in detail with reference to  FIGS. 3 to 5 . 
     The touch controller  43  generates a Tx setup signal for setting a Tx channel to output the driving pulse from the Tx driving circuit  41 , and an Rx setup signal for setting an Rx channel to receive a touch sensor voltage from the Rx driving circuit  42 . In addition, the touch controller  43  generates timing control signals for controlling the operation timings of the Tx driving circuit  41  and the Rx driving circuit  42 . 
     The touch coordinate calculation unit  50  receives the touch raw data TRD from the touch driving circuit  40 . The touch coordinate calculation unit  50  calculates touch coordinates following the calculation method of touch coordinates according to an embodiment, and outputs touch coordinate data including information of the touch coordinates (S 104 ). The touch coordinate calculation unit  50  may be implemented as a micro controller unit (MCU). The host system  70  analyzes the touch coordinate data HIDxy input from the touch coordinate calculation unit  50 , and executes an application program associated with the coordinates at which a touch is generated by a user. 
       FIG. 3  is a block diagram specifically showing an Rx driving circuit in  FIG. 1 .  FIG. 4  is a flowchart showing a method for driving an Rx driving circuit according to an embodiment of the present invention.  FIG. 5  is a waveform diagram showing outputs of a noise filter, an integrator, and a sampling circuit. Hereinafter, the Rx driving circuit  42  and a driving method thereof will be described in detail with reference to  FIGS. 3 to 5 . 
     Referring to  FIG. 3 , the Rx driving circuit  42  includes noise filters NF, integrators INT, sampling circuits SHA, a multiplexer MUX, and an analog to digital converter ADC, which are connected to the respective Rx lines R 1  to Ri. 
     First, the noise filters NF remove noise of signals received from the Rx lines as shown in  FIG. 5 . Specifically, a noise filter NF removes high-frequency noise from a positive signal or a negative signal to output the resultant signal to the integrator INT. In the case where one Rx line of adjacent Rx lines outputs a positive signal, the other Rx line outputs a negative signal. 
     Particularly, the noise filter NF may include variable resistors to completely remove noise of the signals received from the Rx line. In this case, the noise filter NF may change the pass band by adjusting the resistance values of the variable resistors to change the center frequency, as shown in  FIG. 6 . For example, the noise filter NF may change the pass band by adjusting the resistance values of the variable resistors to change the center frequency ωp into any one of first to fifth center frequencies Wp 1  to Wp 5 . The change of the pass band by adjusting the resistance values of the variable resistors of the noise filter N will be later described in detail with reference to  FIGS. 7 to 9  (S 201 ). 
     Second, the integrator INT accumulates the charge variations which pass through the noise filter NF. Specifically, the integrator INT accumulates and adds the positive or negative signal, which passes through the noise filter NF and then is input, P times (P is a natural number), thereby increasing the size of the charge variation.  FIG. 5  illustrates a case where P is 4, but the present invention is not limited thereto. 
     When the high-frequency noise is not removed by the noise filter NF, the high-frequency noise is also accumulated and added by the integrator INT, causing a problem in that the signal to noise ratio (SNR) is decreased (S 202 ). 
     Third, the sampling circuit SHA samples the charge variation accumulated by the integrator INT (S 203 ), as shown in  FIG. 5 . 
     Fourth, the multiplexer MUX receives the charge variation sampled by the respective sampling circuits SHA of the first to (i)th Rx lines R 1  to Ri. The multiplexer MUX sequentially outputs the charge variations, which are sampled by the sampling circuits SHA of the first to i-th Rx lines R 1  to Ri, to the analog to digital converter ADC, by a predetermined control. The analog to digital converter ADC converts the sampled charge variations, which are sequentially input from the multiplexer MUX, into touch raw data TRD, and outputs the converted data (S 204 ). 
       FIG. 7  is a circuit diagram showing a noise filter according to a first embodiment. Referring to  FIG. 7 , a noise filter according to a first embodiment may be implemented as a biquad bandpass filter. 
     The biquad bandpass filter according to a first embodiment is implemented in a fully differential type, and includes two fully differential amplifiers, a plurality of variable resistors, and a plurality of variable capacitors. The biquad bandpass filter may change the center frequency (ωp) by adjusting resistance values of the variable resistors, as shown in  FIG. 6 . In addition, the biquad bandpass filter can reciprocally change a Q-factor by controlling the variable resistors and the variable capacitors. As shown in  FIGS. 8A and 8B , the greater the Q-factor, the narrower the pass band width (bpw), and the smaller the Q-factor, the wider the pass band width (bpw). Resultantly, the present embodiment can change the pass band by adjusting the resistance values of the variable resistors to change the center frequency (Wp), and can change the pass band width (bpw) by adjusting the resistance values of the variable resistors and the capacitance values of the variable capacitors. 
     Now, the biquad bandpass filter according to a first embodiment will be described in detail with reference to  FIG. 7 . The biquad bandpass filter includes first and second input terminals IN 1  and IN 2 , first and second output terminals OUT 1  and OUT 2 , first and second fully differential amplifiers FDA 1  and FDA 2 , a plurality of variable resistors R 11 , R 12 , R 21 , R 22 , R 31 , and R 32 , and a plurality of variable capacitors C 11 , C 12 , C 21 , C 22 , C 31 , and C 32 . 
     The first and second input terminals IN 1  and IN 2  are connected to adjacent Rx lines. That is, in the case where the first input terminal IN 1  is connected to a (k)th Rx line, the second input terminal IN 2  is connected to a (k+1)th Rx line. The reason is that the biquad bandpass filter uses the first and second fully differential amplifiers FAD 1  and FAD 2 , which amplify the difference in charge variation input through the adjacent Rx lines. 
     Each of the first and second fully differential amplifiers FDA 1  and FDA 2  includes a negative polarity input terminal (i(−)), a positive terminal (i(+)), a positive output terminal (o(+)), and a negative output terminal (o(−)). The first output terminal (OUT 1 ) is connected to the positive output terminal (o(+)) of the second fully differential amplifier FDA 2 . The second output terminal (OUT 2 ) is connected to the negative output terminal (o(−)) of the second fully differential amplifier FDA 2 . 
     The plurality of variable resistors includes (1-1)th, (1-2)th, (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 11 , R 12 , R 21 , R 22 , R 31 , and R 32 . The (1-1)th variable resistor R 11  is connected between the negative input terminal (i(−)) and the positive output terminal (o(+)) of the first fully differential amplifier FDA 1 . The (1-2)th variable resistor R 12  is connected between the positive input terminal (i(+)) and the negative output terminal (o(−)) of the first fully differential amplifier FDA 1 . The (2-1)th variable resistor R 21  is connected between the positive output terminal (o(+)) of the first fully differential amplifier FDA 1  and the negative input terminal (i(−)) of the second fully differential amplifier FDA 2 . The (2-2)th variable resistor R 22  is connected between the negative output terminal (o(−)) of the first fully differential amplifier FDA 1  and the positive input terminal (i(+)) of the second fully differential amplifier FDA 2 . The (3-1)th variable resistor R 31  is connected between the negative input terminal (i(−)) of the first fully differential amplifier FDA 1  and the negative output terminal (o(−)) of the second fully differential amplifier FDA 2 . The (3-2)th variable resistor R 32  is connected between the positive input terminal (i(+)) of the first fully differential amplifier FDA 1  and the positive output terminal (o(+)) of the second fully differential amplifier FDA 2 . 
     The plurality of variable capacitors include (1-1)th, (1-2)th, (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable capacitors C 11 , C 12 , C 21 , C 22 , C 31 , and C 32 . The (1-1)th variable capacitor C 11  is connected between the first input terminal IN 1  and the negative input terminal (i(−)) of the first fully differential amplifier FDA 1 . The (1-2)th variable capacitor C 12  is connected between the second input terminal IN 2  and the positive input terminal (i(+)) of the first fully differential amplifier FDA 1 . The (2-1)th variable capacitor C 21  is connected between the negative input terminal (i(−)) and the positive output terminal (o(+)) of the first fully differential amplifier FDA 1 . The (2-2)th variable resistor C 22  is connected between the positive input terminal (i(−)) and the negative output terminal (o(+)) of the first fully differential amplifier FDA 1 . The (3-1)th variable capacitor C 31  is connected between the negative input terminal (i(−)) and the positive output terminal (o(+)) of the second fully differential amplifier FDA 2 . The (3-2)th variable capacitor C 32  is connected between the positive input terminal (i(+)) and the negative output terminal (o(−)) of the second fully differential amplifier FDA 2 . 
     The transfer function of the biquad bandpass filter may be calculated by equation 1. 
     
       
         
           
             
               
                 
                   
                     
                       V 
                       
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                         11 
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                                 21 
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                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     1 
                   
                   ] 
                 
               
             
           
         
       
     
     In equation 1, VI 1  represents the voltage that is input through the first input terminal IN 1  and VO 1  represents the voltage that is output through the first output terminal OUT 1   
     Meanwhile, when the transfer function of equation 1 is expressed by a transfer function using gain value (K), center frequency (ωp), and Q-factor (Q) as variables as shown in equation 2, the center frequency (ωp) and the Q-factor (Q) may be changed by adjusting the resistance values of the variable resistors and the capacitance values of the variable capacitors. In equation 2, s represents the laplace domain. 
     
       
         
           
             
               
                 
                   
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                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     2 
                   
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     In order to express the transfer function of equation 1 by a transfer function using gain value (K), center frequency (ωp), and Q-factor (Q) as variables as shown in equation 2, the (1-2)th variable resistor R 12  and the (1-3)th variable resistor R 13  of equation 1 may be set to have the same resistance value, a first resistance value (RV 1 ), and the (1-2)th variable capacitor C 12  and the (1-3)th variable capacitor C 13  of equation 1 may be set to have the same capacitance value, a first capacitance value (CV 1 ). In this case, the transfer function of the biquad bandpass filter may be calculated by equation 3. 
     
       
         
           
             
               
                 
                   
                     
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                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
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                     3 
                   
                   ] 
                 
               
             
           
         
       
     
     Eventually, when compared equation 2 and equation 3 with each other, the gain value (K) may be calculated as shown in equation 4, the center frequency (ωp) may be calculated as shown in equation 5, and the Q-factor (Q) may be calculated as shown in equation 6. 
     
       
         
           
             
               
                 
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                   [ 
                   
                     Equation 
                     ⁢ 
                     
                         
                     
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                     5 
                   
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                   Q 
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     Therefore, the biquad bandpass filter may change the center frequency (Fc) by adjusting the first resistance value (RV 1 ) corresponding to the resistance values of the (1-2)th variable resistor R 12  and the (1-3)th variable resistor R 13  as shown in  FIGS. 9 a  and 9 b   . For example, as shown in  FIG. 9 a   , the first resistance value (RV 1 ) is adjusted from 120 kΩ to 200 kΩ, thereby changing the center frequency (Fc) from 500 kHz to 300 kHz. 
     In addition, the biquad bandpass filter may change the Q-factor (Q) by adjusting the first resistance value (RV 1 ), and a second resistance value (RV 2 ) corresponding to the resistance values of the (1-1)th variable resistor R 11 , as shown in  FIGS. 9 a  and 9 b   . Particularly, the Q-factor (Q) of the biquad bandpass filter can be reciprocally changed as shown in  FIGS. 9 a  and 9 b   . For example, as shown in  FIG. 9 a   , when the first resistance value (RV 1 ) is adjusted to 120 kΩ and the second resistance value (RV 2 ) is adjusted to 60 kΩ, the Q-factor (Q) becomes 0.5. In addition, as shown in  FIG. 9 b   , when the first resistance value (RV 1 ) is adjusted to 60 kΩ and the second resistance value (RV 2 ) is adjusted to 120 kΩ, the Q-factor (Q) becomes 2. 
     Even through the Q-factor (Q) is changed, the value obtained by multiplying the first resistance value (RV 1 ) by the first capacitance value (CV 1 ) corresponding the capacitance values of the (1-2)th variable capacitor C 12  and the (1-3)th variable capacitor C 13  needs to be constant at the same center frequency (Fc), as shown in  FIGS. 9A and 9B . That is, in the case where the center frequency (Fc) is 500 kHz, the first resistance value (RV 1 ) is 120 kΩ and the first capacitance value (CV 1 ) is 2.65 pF when the Q-factor (Q) is 0.5. In addition, in the case where the center frequency (Fc) is 500 kHz, the first resistance value (RV 1 ) is 60 kΩ and the first capacitance value (CV 1 ) is 5.3 pF when the Q-factor (Q) is 2. Therefore, when the Q-factor (Q) is 0.5 or 2, the values obtained by multiplying the first resistance value (RV 1 ) by the first capacitance value are the same as each other. 
     As above, for the convenience of description, the present invention has been described based on the first input terminal IN 1 , the first output terminal OUT 1 , the (1-1)th, (2-1)th, and (3-1)th variable resistors R 11 , R 21 , and R 31 , and the (1-1)th, (2-1)th, and (3-1)th variable capacitors C 11 , C 21 , and C 31  of the biquad bandpass filter. However, the second input terminal IN 2 , second output terminal OUT 2 , the (1-2)th, (2-2)th, and (3-2)th variable resistors R 11 , R 21 , and R 31 , and the (1-2)th, (2-2)th, and (3-2)th variable capacitors C 11 , C 21 , and C 31  of the biquad bandpass filter are substantially the same as those as described above. 
     As described above, the present embodiments can change the pass band by adjusting the resistance values of the variable resistors of the biquad bandpass filter to change the center frequency, and can change the pass band width by adjusting the resistance values of the variable resistors and the capacitance values of the capacitance values of the variable capacitors. 
       FIGS. 10A and 10B  are exemplary views specifically showing variable resistors and variable capacitors. Referring to  FIG. 10A , each of the variable resistors R 11 , R 12 , R 21 , R 22 , R 31 , and R 32  of the biquad bandpass filter includes a first resistor R 1 , a second resistor R 2 , and switches SW 1  and SW 2 . 
     The first resistor R 1  and the second resistor R 2  are connected in parallel. The resistance value of the first resistor R 1  may be substantially the same as the resistance value of the second resistor R 2 . Both ends of the second resistor R 2  are connected to the switches SW 1  and SW 2 . That is, one end of the second resistor R 2  is connected to the first switch SW 1 , and the other end of the second resistor R 2  is connected to the second switch SW 2 . 
     The switches SW 1  and SW 2  are turned on in response to a predetermined control signal. For example, the switches SW 1  and SW 2  may be controlled by a control signal which swings between a first voltage and a second voltage, and may be designed so as to be turned on when a control signal of the first voltage is input and turned off when a control signal of the second voltage is input. 
     In addition, when the switches SW 1  and SW 2  are turned on, the variable resistors are connected in parallel, and thus the resistance value of the variable resistors when the switches SW 1  and SW 2  are turned on is ½ times the resistance value of the variable resistors when the switches SW 1  and SW 2  are turned off. In this case, the present invention can reciprocally change the Q-factor (Q) by adjusting the resistance values of the variable resistors and the capacitance values of the variable capacitors. Specifically, the resistance value of the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32  and the resistance value of the respective (1-1)th and (1-2)th variable resistors R 11  and R 12  can be reciprocally changed by oppositely controlling the control signal supplied to the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32  and the control signals supplied to the respective (1-1)th and (1-2)th variable resistors R 11  and R 12 . 
     For example, when the control signal having the first voltage is supplied to the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32 , the control signal having the second voltage is supplied to the respective (1-1)th and (1-2)th variable resistors R 11  and R 12 , so that the resistance value of the (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32  can be controlled to be ½ times the resistance value of the (1-1)th and (1-2)th variable resistors R 11  and R 12 . Alternatively, when the control signal having the second voltage is supplied to the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32 , the control signal having the first voltage is supplied to the respective (1-1)th and (1-2)th variable resistors R 11  and R 12 , so that the resistance value of the (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable resistors R 21 , R 22 , R 31 , and R 32  can be controlled to be 2 times the resistance value of the (1-1)th and (1-2)th variable resistors R 11  and R 12 . 
     Referring to  FIG. 10 b   , each of the variable capacitors C 11 , C 12 , C 21 , C 22 , C 31 , and C 32  of the biquad bandpass filter includes a first capacitor C 1 , a second capacitor C 2 , and switches SW 3  and SW 4 . 
     The first capacitor C 1  and the second capacitor C 2  are connected in parallel. The capacitance value of the first capacitor C 1  may be substantially the same as the capacitance value of the second capacitor C 2 . Both ends of the second capacitor C 2  are connected to the switches SW 3  and SW 4 . That is, one end of the second capacitor C 2  is connected to the third switch SW 1  and the other end of the second capacitor C 2  is connected to the fourth switch SW 4 . 
     The switches SW 3  and SW 4  are turned on in response to a predetermined control signal. For example, the switches SW 3  and SW 4  may be controlled by a control signal which swings between a first voltage and a second voltage, and may be designed so as to be turned-on when a control signal of the first voltage is input and turned-off when a control signal of the second voltage is input. 
     In addition, when the switches SW 3  and SW 4  are turned on, the capacitance value of the variable capacitors when the switches SW 3  and SW 4  are turned on is 2 times the capacitance value of the variable capacitors when the switches SW 1  and SW 2  are turned off since the variable capacitors are connected in parallel. Resultantly, the resistance value of the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable capacitors C 21 , C 22 , C 31 , and C 32  and the capacitance value of the respective (1-1)th and (1-2)th variable capacitors C 11  and C 12  can be reciprocally changed by oppositely controlling the control signal supplied to the respective (2-1)th, (2-2)th, (3-1)th, and (3-2)th variable capacitors C 21 , C 22 , C 31 , and C 32  and the control signal supplied to the respective (1-1)th and (1-2)th variable capacitors C 11  and C 12 . 
       FIG. 11  is a circuit diagram showing a noise filter according to a second embodiment. Referring to  FIG. 11 , a noise filter according to the second embodiment may be implemented as a biquad bandpass filter. 
     The biquad bandpass filter according to the second embodiment is implemented in a single-ended type, and includes two differential amplifiers, a plurality of variable resistors, a plurality of variable capacitors, and an inverting amplifier. The biquad bandpass filter may change the center frequency (ωp) by adjusting resistance values of variable resistors, as shown in  FIG. 6 . In addition, the biquad bandpass filter can reciprocally change the Q-factor by controlling the variable resistors and the variable capacitors. As shown in  FIG. 8 , the greater the Q-factor, the narrower the pass band width (bpw), and the smaller the Q-factor, the wider the pass band width (bpw). Resultantly, the present invention can change the pass band by adjusting the resistance values of the variable resistors to change the center frequency (ωp), and can change the pass band width (bpw) by adjusting the resistance values of the variable resistors and the capacitance values of the variable capacitors. 
     Now, the biquad bandpass filter according to the second embodiment will be described in detail with reference to  FIG. 11 . The biquad bandpass filter includes a first input terminal IN 1 , a first output terminal OUT 1 , first and second differential amplifiers DA 1  and DA 2 , a plurality of variable resistors R 11 , R 21 , and R 31 , a plurality of capacitors C 11 , C 21 , and C 31 , and an inverting amplifier IA. 
     The first input terminal IN 1  is connected to the Rx line. Each of the first and second differential amplifiers DA 1  and DA 2  includes a negative input terminal (−), a positive input terminal (+), and an output terminal (o). The first output terminal (OUT 1 ) is connected to the output terminal (o) of the second differential amplifier DA 2 . 
     The plurality variable resistors include first, second, and third variable resistors R 11 , R 21 , and R 31 . The first variable resistor R 11  is connected between the negative input terminal ((−) and the positive output terminal (o(+)) of the first differential amplifier DA 1 . The second variable resistor R 21  is connected between the output terminal (o) of the first differential amplifier DA 1  and the negative input terminal (−) of the second differential amplifier DA 2 . The third variable resistor R 31  is connected between the negative input terminal (−) of the first differential amplifier DA 1  and the output terminal (o) of the second differential amplifier DA 2 . 
     The plurality variable capacitors include first, second, and third variable capacitors C 11 , C 21 , and C 31 . The first capacitor C 11  is connected between the first input terminal IN 1  and the negative input terminal (−) of the first differential amplifier DA 1 . The second variable capacitor C 21  is connected between the negative input terminal (−) and the output terminal (o(+)) of the first differential amplifier DA 1 . The third variable capacitor C 31  is connected between the negative input terminal (−) and the output terminal (o) of the second differential amplifier DA 2 . 
     The inverting amplifier IA is connected between the first output terminal OUT 1  and the output terminal (o) of the second differential amplifier DA 2 . 
     The transfer function of the biquad bandpass filter according to the second embodiment may be calculated by equation 2. Here, when the second and third variable resistors R 2  and R 3  are set to have the same resistance value, a first resistance value RV 1 , and the (1-2)th variable capacitor C 12  and the (1-3)th variable capacitor C 13  are set to have the same capacitance value, a first capacitance value CV 1 , the transfer function of the biquad bandpass filter according to the second embodiment may be calculated by equation 3. Therefore, the gain value (K) may be calculated as shown in equation 4, the center frequency (Wp) may be calculated as shown in equation 5, and the Q-factor (Q) may be calculated as shown in equation 6. 
     In addition, the respective variable resistors R 11 , R 21 , and R 31  of the biquad bandpass filter according to the second embodiment may be substantially the same as those as described with reference to  FIG. 10A , and the respective variable capacitors C 11 , C 21 , and C 31  may be substantially the same as those as described with reference to  FIG. 10B . 
     The charge variation and the noise of each touch sensor may be different when the touch screen panel TSP is touched by a part of user&#39;s body and when the touch screen panel TSP is touched by a pen. Due to this, when the noise filter is set to be suitable for the case where the touch screen panel TSP is touched by a part of user&#39;s body, the noise may not be removed when the touch screen panel TSP is touched by a pen. In order to prevent this, the present embodiment may be configured such that the case where the touch screen panel TSP is touched by a part of user&#39;s body and the case where the touch screen panel TSP is touched by a pen are sensed differentially, and the pass band and the pass band width of the biquad bandpass filter are changed when the touch screen panel TSP is touched by a part of user&#39;s body and when the touch screen panel TSP is touched by a pen. As a result, the present embodiment can minimize the noise of the charge variation of the respective touch sensors. 
     As described above, the present invention can change the pass band by implementing the noise filter as the biquad bandpass filter and adjusting the resistance values of the variable resistors of the biquad bandpass filter to change the center frequency, and can change the pass band width by adjusting the resistance values of the variable resistors and the capacitance values of the variable capacitors. As a result, the present invention can change the pass band and the pass band width in advance so as to be optimized for removing the noise of the charge variation of each of the touch sensors before product shipping, thereby minimizing the noise of the charge variations of the respective touch sensors. 
     Further, the present invention can differentially sense the case where the touch screen panel TSP is touched by a part of user&#39;s body and the case where the touch screen panel TSP is touched by a pen, and change the pass band and the pass band width of the biquad bandpass filter when the touch screen panel TSP is touched by a part of user&#39;s body and when the touch screen panel TSP is touched by a pen. As a result, the present invention can minimize the noise of the charge variation of each of the touch sensors. 
     Further, the present invention can prevent the overflow of the charge variations accumulated in the integrator by minimizing the noise of the charge variation of each of the touch sensors to reduce the size of the charge variation accumulated in the integrator. 
     Further, the present invention can increase the number of times of integration of the charge variation accumulated in the integrator by minimizing the noise of the charge variation of each of the touch sensors to reduce the sizes of the charge variation accumulated in the integrator. As a result, the present invention can further improve the accuracy in the touch coordinate calculation. 
     Further, the present invention can improve the signal to noise ratio (SNR) by minimizing the noise of the charge variation of each of the touch sensors. 
     Although embodiments have been described with reference to a number of illustrative embodiments thereof, it should be understood that numerous other modifications and embodiments can be devised by those skilled in the art that will fall within the scope of the principles of this disclosure. More particularly, various variations and modifications are possible in the component parts and/or arrangements of the subject combination arrangement within the scope of the disclosure, the drawings and the appended claims. In addition to variations and modifications in the component parts and/or arrangements, alternative uses will also be apparent to those skilled in the art.