Patent Publication Number: US-2023138691-A1

Title: Ratiometric sensor circuit

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     This application claims priority to German Patent Application No. 102021128249.7 filed on Oct. 29, 2021, the content of which is incorporated by reference herein in its entirety. 
     TECHNICAL FIELD 
     The present disclosure relates generally to a sensor circuit and in particular to sensor circuits having a ratiometric behavior. 
     BACKGROUND 
     In electronics the term “ratiometric” means that an unknown variable can be derived from a known ratio of a plurality of other variables to one another. Generally, during a ratiometric measurement as a quotient to two variables with the same superposed disturbance, it turns out that the latter does not influence the measurement. A ratiometric measurement variable (such as e.g. a measured magnetic field strength) is for example independent of a supply voltage (V DD ) that may be subject to fluctuations. If the supply voltage rises unexpectedly at a measurement system, for example, a measurement signal coupled linearly to the supply voltage thus rises concomitantly. The variable to be measured has not changed, however. If the measurement signal were then passed to an ADC (Analog-to-converter) with a fixed reference voltage, the result would be that the ADC supplies a code that corresponds to a higher measurement variable—incorrect measurement—not a ratiometric system in this case. If the reference voltage of the ADC also concomitantly rose linearly with respect to the supply, the variable to be measured would not change at the output of the ADC—correct measurement—since a ratiometric system is present in this case. Ratiometric therefore means that if the measured signal changes as a result of a disturbance variable in the system, then the comparison variable must change in the same way, such that the change as it were “cancels out” and the change remains with “1/1”. The measurement signal is then multiplied by this fraction. The ratiometric is perfect with 1, already poorer with 1.15, etc. 
     Sensor circuits having a fully ratiometric output (ratiometric sensor circuits) can be realized using digital signal processing, for example, but are relatively more expensive and slow. Ratiometric sensor circuits can for example also be realized using digitally assisted amplifiers having a digitally controlled gain. These do not have a problem with regard to speed, but are even more expensive. Sensors having a ratiometric quiescent voltage but a fixed sensitivity (=gain) avoid an analog multiplication by the supply voltage, but do not offer the functionality of a mechanical potentiometer and are not appropriate for a low-cost microprocessor (which receives the sensor signal) which uses the supply voltage as ADC reference (ADC=analog-to-digital converter). A sensor having a fixed sensitivity (gain) does not offer a complete ratiometric reaction. 
     Therefore, there is a need for improved ratiometric sensor circuits with at the same time manageable costs. 
     SUMMARY 
     This need is met by circuits and methods as claimed in the independent claims. The dependent claims relate to advantageous developments. 
     In accordance with a first aspect of the present disclosure, a sensor circuit is proposed which has a control circuit configured to control a constant first signal (e.g. constant current or constant voltage) to a ratiometric second signal (e.g. ratiometric current or ratiometric voltage) using a first amplifier adjustable by an actuating or control signal. The second signal is thus linearly coupled to a supply signal (e.g. supply voltage). The sensor circuit includes an adjustable second amplifier for a sensor signal, the gain of which is adjustable by the control signal of the control circuit. As a result, a ratiometric gain can be obtained at the second amplifier for the sensor signal. As a result, the gain of the second amplifier is likewise linearly coupled to the supply signal (e.g. supply voltage). With the proposed circuit arrangement, a ratiometric behavior can be attained with relatively simple and cost-effective amplifier designs. 
     In accordance with some example implementations, the first and second amplifiers are configured as replica amplifiers, e.g. as copies of one another. If for example the second amplifier for the sensor signal is regarded as the main amplifier, then the first amplifier can be configured as an exact copy of the main amplifier. However, the first amplifier of the control circuit does not amplify the sensor signal, but rather serves for adjusting the control signal for the (structurally identical) second amplifier. The term “amplifier” here does not necessarily mean just a single amplifier element (such as e.g. a transistor), but rather can mean a complete amplifier circuit which can have a plurality of circuit components, such as e.g. bipolar or MOS differential stages, cascodes, etc. The purpose behind using a replica amplifier may be that both amplifiers always behave identically even under varying ambient conditions. “Replica amplifiers” can be understood to mean not only amplifier circuits that are identical (within the scope of manufacturing tolerances), but also so-called scalable replica amplifiers. In the case of the latter, for example, a gain factor can differ between the two amplifiers (e.g. as a result of different emitter areas or source-drain channels), with an otherwise identical implementation. 
     In accordance with example implementations, the control circuit with the replica amplifier (replication amplifier) thus supplies a control signal which controls the gain of the sensor signal, such that an (amplified) output signal proportional to a supply voltage (ratiometric sensitivity=ratiometric gain) can be obtained. Technology-dependent fluctuations and temperature fluctuations of imprecise amplifiers can thus be compensated for. 
     In accordance with some example implementations, the sensor circuit is an integrated sensor circuit and the first and second amplifiers are formed on a common semiconductor substrate (or chip, die). Very exact replica circuits of main circuits can be produced in this way since both circuits are subject to the same production processes and production conditions. 
     In accordance with some example implementations, the sensor circuit includes a sensor configured for outputting the sensor signal in response to a physical measurement variable. The sensor can be embodied for example on the same semiconductor substrate as the other circuit components (e.g. first and second amplifiers) of the sensor circuit. 
     In accordance with some example implementations, the sensor includes a magnetic field sensor, such as, for example, a Hall sensor or a magnetoresistive sensor (xMR sensor). The Hall sensor can be configured for spinning current operation, for example. A Hall sensor includes one or more Hall effect sensor elements that measure a magnetic field strength and/or direction. These measurements are used to obtain parameters, such as, for example, distance, position and rotational speed. However, Hall sensor elements exhibit offsets at their outputs on account of mechanical loads, doping and geometric defects. Furthermore, Hall sensor elements exhibit an offset drift that results in an unpredictable and temporally variable output error. Such offsets in the case of Hall sensor elements can be reduced using the spinning current method, in which the bias current of a Hall sensor element is rotated spatially around the Hall sensor element, while the output signals are averaged over time. This can reduce an offset and an offset drift. 
     In accordance with some example implementations, the sensor circuit includes a reference voltage source configured to provide a reference voltage as the constant first signal. The reference voltage source can include a bandgap reference, for example. The term bandgap reference denotes a reference voltage source whose output voltage in a temperature-compensated state corresponds to the bandgap voltage of a semiconductor. The voltage generated thus varies depending on the semiconductor material. One particular property of a bandgap reference is a high precision in conjunction with low outlay in terms of circuitry. Moreover, bandgap references exhibit temperature stability and have a low terminal voltage (&lt;3 volts). 
     In accordance with some example implementations, the sensor circuit includes a voltage divider connected between an external supply voltage and ground and configured to provide a ratiometric voltage as the second signal. The voltage divider is linearly coupled to the supply voltage, which may be subject to fluctuations. The voltage divider can be formed by electrical resistors, such that a voltage linearly coupled to the supply voltage is thereby obtained as a controlled variable for the control circuit. 
     In accordance with some example implementations, the first and second amplifiers each include a MOS differential stage (MOS=metal oxide semiconductor) having a first MOSFET and second MOSFET (MOSFET=Metal Oxide Semiconductor Field Effect Transistor). The source and drain currents of the MOSFETs are adjustable in each case by the control circuit. In predefined input and output voltages of the first (replica) amplifier, it is thus possible to adjust the gain thereof between input voltage and output voltage. It goes without saying that bipolar differential stages could also be used. Optionally, (folded) cascode structures can also be coupled to the outputs of the differential stages. 
     In accordance with some example implementations, the control circuit includes a transconductance amplifier having a first input for a first signal amplified using the first amplifier and a second input for the ratiometric second signal (e.g. voltage at a voltage divider), wherein the transconductance amplifier is configured to output the actuating signal at its output depending on a difference between the inputs. A transconductance amplifier is a special operational amplifier that converts the differential voltage at the two inputs into a proportional output current. Like other operational amplifiers it can usually be realized as an integrated circuit. Controlled current sources can then be driven via the output of the transconductance amplifier in order to adjust source and drain bias currents of the MOSFETs of the differential stages. 
     In accordance with some example implementations, the control circuit includes a comparator having a first input for a first signal amplified using the first amplifier and a second input for the ratiometric second signal and a charge pump coupled to an output of the comparator, wherein the comparator is configured to output the actuating signal at its output depending on a difference between the inputs. Using the actuating signal it is possible to drive the charge pump and via the latter in turn controlled current sources in order to adjust source and drain bias currents of the MOSFETs of the differential stages. 
     In accordance with some example implementations, the first and second amplifiers are each configured as a chopper amplifier. Chopper amplifiers are a type of amplifier in which a signal to be amplified is modulated (chopped), amplified and demodulated again. Use of such a technology makes it possible to shift a zero error (or offset error) and so-called 1/f noise of an amplifier into a frequency band which is not of interest. Such chopper amplifiers can be used for example in bandgap circuits which provide a defined reference voltage, but can likewise be used in other applications in which a signal is to be amplified, such as, for example, for amplifying measurement signals. 
     In accordance with some example implementations, the first and/or the second (chopper) amplifier can each have at least one capacitively coupled chopper demodulator in order to further smooth a chopper ripple. In the case of a capacitively coupled chopper demodulator, an output of an amplifier can be coupled to a downstream amplifier stage via one or more capacitances in order thereby to smooth a chopper ripple. In this case, the capacitances can be switched directly into the signal path between output of the amplifier and input of the downstream amplifier stage and/or between the signal path and ground. 
     In accordance with some example implementations, a feedback loop having switched capacitors can be coupled to an output at least of the second amplifier (main amplifier), and feeds back a chopper ripple. A chopper ripple can be smoothed as a result. By virtue of a feedback loop having switched capacitors, the discharge resistances can be realized by switched capacitances (referred to as switched capacitors). The term switched capacitor filters, or often also just SC filters for short, denotes electronic filters in which ohmic resistors are replaced by switched capacitors. They are time-discrete filters. The filter parameters of the SC filters can be altered very easily by varying the switching frequency f s  with which the capacitances are switched over. The replacement of the ohmic resistances R in a given circuit, such as a low-pass filter, by capacitances C S  which are operated with the switchover frequency f s  can be calculated in accordance with R=1/f s C S . It will be apparent to the person skilled in the art that the switching frequency f s  of the capacitances need not correspond to the chopper frequency f chop . 
     In accordance with a further aspect of the present disclosure, a method for operating a sensor circuit is proposed, including controlling a constant first signal to a ratiometric second signal using a first amplifier adjustable by an actuating or control signal, and adjusting a gain of an adjustable second amplifier for a sensor signal based on the actuating signal. 
     In accordance with some example implementations, the first and second amplifiers are operated as identical or scaled replica amplifiers. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Some examples of devices and/or methods are explained in greater detail; merely by way of example below with reference to the accompanying figures, in which: 
         FIG.  1    shows a block diagram of a ratiometric sensor circuit in accordance with one example implementation; 
         FIG.  2    shows a further block diagram of a ratiometric sensor circuit in accordance with one example implementation; 
         FIG.  3    shows one example implementation of an integrated ratiometric sensor circuit comprising MOS differential stages; 
         FIG.  4    shows a further example implementation of an integrated ratiometric sensor circuit comprising chopper amplifiers; 
         FIG.  5    shows a further example implementation of an integrated ratiometric sensor circuit comprising chopper amplifiers; and 
         FIG.  6    shows a further example implementation of an integrated ratiometric sensor circuit comprising chopper amplifiers and a feedback loop having switched capacitors for suppressing the chopper ripple. 
     
    
    
     DETAILED DESCRIPTION 
     Some examples will now be described more thoroughly with reference to the accompanying figures. However, further possible examples are not restricted to the features of these implementations described in detail. These may have modifications of the features and counterparts and alternatives to the features. Furthermore, the terminology used herein for describing specific examples is not intended to be limiting for further possible examples. 
     Throughout the description of the figures, identical or similar reference signs refer to identical or similar elements or features which can be implemented in each case identically or else in modified form, while they provide the same or a similar function. In the figures, furthermore, the thicknesses of lines, layers and/or regions may be exaggerated for the purpose of clarity. 
     If two elements A and B are combined using an “or” this should be understood such that all possible combinations are disclosed, e.g. only A, only B and A and B, unless expressly defined otherwise in an individual case. As alternative wording for the same combinations, it is possible to use “at least one from A and B” or “A and/or B”. That applies equivalently to combinations of more than two elements. 
     If a singular form, e.g. “a”, “an” and “the”, is used and the use of only a single element is defined neither explicitly nor implicitly as obligatory, further examples can also use a plurality of elements in order to implement the same function. If a function is described below as being implemented using a plurality of elements, further example can implement the same function using a single element or a single processing entity. Furthermore, it goes without saying that the terms “comprises”, “comprising”, “has” and/or “having” in their usage describe the presence of the specified features, integers, steps, operations, processors, elements, components and/or a group thereof, but do not exclude the presence or the addition of one or more other features, integers, steps, operations, processors, elements, component and/or a group thereof. 
     The concept of a ratiometric sensor circuit proposed herein has, in principle, numerous possibilities for use in various electrical and/or electronic circuits. Therefore, the circuit concept disclosed herein is initially explained very generally with reference to  FIGS.  1  and  2   . 
       FIG.  1    shows a basic illustration of a sensor circuit  100  in accordance with one example implementation. 
     The sensor circuit  100  comprises a control circuit  110  configured to control a constant first signal  112 , such as e.g. a constant voltage, to a ratiometric second signal  116 , such as e.g. a ratiometric voltage, using a first amplifier (not shown) adjustable by an actuating or controlled signal  114 . The sensor circuit  100  additionally comprises an adjustable second amplifier  120  for a sensor signal  132 , the gain of which is adjustable by the actuating signal  114  of the control circuit  110 . 
     The control circuit  110  thus controls the gain V 1  of the first amplifier (not shown) in accordance with a ratio between the ratiometric second signal  116  (V ratio ) and the constant first signal  112  (V const ), e.g. V 1 =V ratio /V const . In this case, the ratiometric second signal  116  is linearly dependent on a supply signal  140 , such as e.g. an external supply voltage V DD,ext . The supply signal  140  can be subject to fluctuations and thus for example also ratiometrically influence the sensor signal  132 . The ratiometric second signal  116  can correspond to the supply voltage V DD,ext    140  itself, or a signal dependent thereon. In the example shown in  FIG.  1   , the sensor circuit comprises a voltage divider  150  connected between the external supply voltage  140  and ground and configured to provide the ratiometric voltage V ratio  as the second signal  116 , to which the control circuit  110  effects control. The voltage divider  150  here comprises by way of example three electrical resistors connected in series, the ratiometric voltage V ratio  being tapped off at terminals of a middle resistor. It goes without saying that the voltage divider  150  could also be configured differently, for example with only two or more than three resistors. 
     The sensor circuit  100  shown in  FIG.  1    comprises a reference voltage source  160  configured to provide the control circuit  110  with a constant reference voltage V const  as the first signal  112 . For this purpose, the reference voltage source  160  can comprise a bandgap reference  162 , which in the example shown is connected between the supply signal  140  and ground. A further voltage divider  170  can be provided between an output of the bandgap reference  162  and ground, and is configured to provide the constant reference voltage V const  as the second signal  116  based on the bandgap voltage at the output of the bandgap reference  162 . It goes without saying that, in other implementations, the bandgap voltage of the bandgap reference  162  could also be used directly as the first signal  112  and the voltage divider  170  could thus be obviated. The voltage divider  170  here comprises by way of example three electrical resistors connected in series, the constant reference voltage V const  being tapped off at terminals of a middle resistor of the voltage divider  170 . It goes without saying that the voltage divider  170  could also be configured differently, for example with only two or more than three resistors. 
     The actuating or control signal  114  for adjusting the gain of the first amplifier (not shown in  FIG.  1   ) is equally fed to the second amplifier  120  for amplifying the sensor signal  132 , such that the gain V 2  thereof likewise behaves in a ratiometric manner (e.g. in a linearly dependent manner with respect to the supply signal  140 ). With a ratiometric sensor signal and a ratiometric gain V 2 , a fully ratiometric system is thus obtained. As sensor  130 ,  FIG.  1    shows by way of example a Hall sensor for measuring a magnetic field. A Hall voltage which is caused by the magnetic field and which behaves in a ratiometric manner is amplified using the second amplifier to form an output signal  122 . Since the gain V 2  of the second amplifier  120  is likewise ratiometric, a fully ratiometric measurement system is obtained. It goes without saying that the present disclosure is not restricted to Hall sensors, but rather is applicable to any sensors. In this regard, for example, other magnetic field sensors, such as e.g. magnetoresistive sensors (AMR, GMR, TMR), or capacitive sensors, inductive sensors, etc., could be used as sensors  130 . In this case, the sensors  130  can have for example a ratiometric quiescent voltage which is linearly dependent on the signal  140 . 
     The first amplifier of the control circuit  110  and the second amplifier  120  for the sensor signal  132  can be configured as identical or scalable replica amplifiers. That is indicated in  FIG.  2   , which schematically illustrates the first amplifier  118  in the control loop  110 . The first amplifier  118  is configured as a replica (copy) of the second amplifier  120 , or vice versa. That means that both amplifiers  118 ,  120  are constructed substantially identically and thus also behave correspondingly identically. That means that for both amplifiers  118 ,  120  the actuating signal  114  leads to the same (ratiometric) gain (V 1 =V 2 ). For this purpose, it is possible to implement the two amplifiers  118 ,  120  with identical circuit structures jointly on a semiconductor chip. A virtually identical behavior can thus be ensured. Besides the two amplifiers  118 ,  120 , it is also possible to integrate even more or all circuit components of the sensor circuit  100  as an integrated circuit jointly on the semiconductor chip. 
     One example implementation of the sensor circuit  100  is shown in  FIG.  3   . 
     The integrated sensor circuit  300  in  FIG.  3    has a left circuit portion and a right circuit portion on a common semiconductor substrate (die). The left circuit portion comprises the control circuit  110  with replica amplifier  118 , and the right circuit portion comprises Hall sensor  130  together with main amplifier  120  for the sensor signal  132 . Replica amplifier  118  and main amplifier  120  are embodied identically (within the scope of manufacturing tolerances), that is to say have identical transistors, for example. Replica amplifier  118  and main amplifier  120  each have a MOS differential stage  302  having in each case a first MOSFET  304  and a second MOSFET  306  and having source and drain bias currents of the MOSFETs  304 ,  306  that are adjustable in each case by the control circuit  110 . For this purpose, a current source  310  controllable using the control circuit  110  is coupled to the integrated source terminals of the MOSFETs  304 ,  306 . Furthermore, a current source  312  controllable using the control circuit  110  is coupled to the drain terminal of the MOSFET  304  and a current source  312  controllable using the control circuit  110  is coupled to the drain terminal of the MOSFET  306 . The MOS differential stages  302  respectively form differential amplifiers  118  and  120  having a respective differential input at the gates of the MOSFETs  304 ,  306  and a respective differential output at the drain terminals of the MOSFETs  304 ,  306 . 
     In the example shown in  FIG.  3   , the control circuit  110  has a transconductance amplifier  308  having a first (differential) input for a (differential) output signal of the replica amplifier  118  and having a second (differential) input for the (differential) ratiometric setpoint signal  116  from the voltage divider  150  coupled between VDD ext  and ground. The differential output signal amplified by the replica amplifier  118  can be tapped off between the drain terminals of the PMOSFETs  304 ,  306 . The transconductance amplifier  308  is configured to output at its output the actuating signal (control current)  114  depending on the difference between the (differential) transconductance amplifier inputs. 
     Using the actuating signal (control current)  114  and one or more current mirrors arranged between the left and right circuit portions, the respective controlled current sources  310 ,  312  can be driven in both circuit portions, the current sources adjusting source and drain currents of the respective MOSFETs  304 ,  306  of the MOS differential stages  302  such that a difference between the (differential) inputs of the transconductance amplifier  308  becomes (substantially) zero. That in turn then means that the output voltage of the replica amplifier  118  adjusted by the control circuit  110  corresponds to the ratiometric setpoint voltage  116 . The controlled current sources  310  are each coupled between a supply potential  320  and source terminals of the respective MOSFETs  304 ,  306  of the amplifiers  118 ,  120 . The gate terminals of the MOSFETs  304 ,  306  of the replica amplifier  118  form a differential input for the constant reference voltage  112 . The gate terminals of the MOSFETs  304 ,  306  of the main amplifier  120  form a differential input for the differential sensor signal  132 . Controlled current sources  312  are in each case coupled between the drain terminals of the respective MOSFETs  304 ,  306  and ground, and are coupled to the actuating signal (control current)  114  via a current mirror. Consequently, both amplifiers  118 ,  120  are ratiometrically controlled identically by way of the control circuit  110 . 
     Optionally, a common-mode output signal can be obtained between the drain terminals of the MOSFETs  304 ,  306  of the respective MOS differential stages  302 . For this purpose, a voltage divider having two identical resistors  314  can be connected between the drain terminals of the respective MOSFETs  304 ,  306 . The resistors  314  can be embodied as silicided polysilicon resistors (poly-resistors), for example. It is assumed here that the integrated sensor circuit  300  or at least parts thereof are embodied using complementary metal oxide semiconductor technology (CMOS). Integrated circuits (ICs) often require integrated resistors for proper operation of the circuit. Such resistors generally consist of doped polycrystalline silicon. In order to reduce the resistances in the poly-resistors, a metal silicide layer can be formed over the doped polycrystalline silicon or the formation of such a layer over the doped polycrystalline silicon can be prevented. This option of the metal silicide layer enables two different types of resistors composed of polycrystalline silicon. The first type with the metal silicide layer over the doped polycrystalline silicon is referred to as “silicided polycrystalline silicon resistor” (silicided poly-resistor), and the electrical conduction of this resistor is effected via the metal silicide layer. The second type without the metal silicide layer over the doped polycrystalline silicon is referred to as “non-silicided polycrystalline resistor” (non-silicided poly-resistor). In the case of the second type, the electrical conduction is effected through the polycrystalline silicon, which depends on the doping (p- or n-doped) of the polycrystalline silicon. The temperature coefficient of the resistance of silicided polysilicon resistors is similar to that of metal resistors. 
     Although  FIG.  3    shows the integrated sensor circuit  300  comprising MOS differential stages  302 , it goes without saying that other implementations of this concept are also possible, such as using bipolar differential stages, for example, in which analogously respective emitter and collector bias currents for adjusting a gain can then be controlled using the control circuit  110 . 
     A further example implementation of the sensor circuit  100  is shown in  FIG.  4   . 
     The integrated sensor circuit  400  in  FIG.  4    differs from the integrated sensor circuit  300  in  FIG.  3    for example in that the control circuit  110  in the left circuit portion comprises, instead of the transconductance amplifier  308 , a comparator  408  having a first (differential) input for a (differential) output signal amplified using a replica amplifier  118  and having a second (differential) input for the (differential) ratiometric setpoint signal  116  from the voltage divider  150  coupled between VDD ext  and ground. An output signal of the comparator  408  acts as an actuating signal  114  and controls a charge pump  410  coupled to the output of the comparator  408 . By way of the charge pump  410 , the controlled current sources  310 ,  312  can in turn be driven, and adjust source and drain currents of the respective MOSFETs  304 ,  306  of the MOS differential stages  302  such that the difference between the (differential) inputs of the comparator  408  becomes zero. It goes without saying that comparator  408  and charge pump  410  could also be used in the example implementation in  FIG.  3   . 
     As a further difference with respect to  FIG.  3   , in the example implementation in  FIG.  4   , both replica amplifiers  118 ,  120  are operated as so-called chopper amplifiers.  FIG.  4    thus shows a control loop  110  for chopped replica amplifiers  118  and chopped sensor amplifier  120  for the compensation of a dynamic gain reducing effect of chopped amplifiers and with comparator  408  and charge pump  410  in the control loop  110  for providing a control signal  114  for current sources  310 ,  312  (which controls the gain of the amplifiers). Both amplifier circuits  118 ,  120  each comprise a modulator circuit  412  at the input of the respective amplifier, the modulator circuit being switched with a chopper frequency f chop  and being configured to convert a respective DC input voltage (DC=Direct Current) into an AC input voltage (AC=Alternating Current), e.g. an AC voltage, in accordance with the chopper frequency. In the case of the amplifier  118 , the input voltage is the constant voltage V const . In the case of the amplifier  120 , the input voltage is the Hall voltage  132 . Both amplifier circuits  118 ,  120  each furthermore comprise, at the output of the respective modulator circuit  412 , the MOS differential stage  302  having a differential input for the AC input voltage and having a differential output (drain terminals of the MOSFETs  304 ,  306 ) for an amplified AC voltage. Both amplifier circuits  118 ,  120  furthermore comprise, at the output of the MOS differential stage  302 , a demodulator circuit  414 , switched with the chopper frequency f chop  and configured to convert the amplified AC voltage back into an amplified DC voltage. 
     The respective demodulated circuits  414  can be configured during different switching phases to couple each of the inverting and noninverting outputs of the amplifier circuit  118  and/or  120  capacitively (e.g. using capacitances) to each inverting and noninverting input of a differential amplifier  408  or  418  connected downstream. For this purpose, capacitances can be provided directly in the respective signal paths and/or be connected between the respective signal paths and ground. Such capacitively coupled demodulated circuits  414  enable a chopper ripple present at the output of the chopper amplifier circuit  118 ,  120  to be reduced further by comparison with conventional (non-capacitively coupled) chopper amplifier circuits. The term chopper ripple denotes offset voltages that are amplified by the amplifier circuit and demodulated by the demodulator circuit  414 . A conventional demodulation of an offset DC voltage then results in an undesired AC voltage (chopper ripple). The proposed capacitively coupled implementation and functioning of the demodulator circuits  414  can reduce the chopper ripple. Capacitively coupled chopper amplifier circuits are described for example in Qinwen Fan, Kofi A. A. Makinwa, Johan H. Huij sing: “Capacitively-Coupled Chopper Amplifiers”, Springer International Publishing, 2017.! 
     A capacitively coupled demodulator circuit  414  can be of interest in particular at the output of the main or measurement amplifier  120  for the sensor signal  132  of the Hall sensor  130 . In the example implementation illustrated, the Hall sensor  130  is operated in a so-called spinning current mode with a spinning frequency f s . During a first switching phase of the Hall sensor  130 , a first current flows via two first out of four terminals of the Hall sensor  130 , such that a first Hall voltage can be tapped off at the other two terminals. During a succeeding second switching phase, a second current flows via the other two terminals, such that a second Hall voltage can be tapped off at the first terminals, etc. It goes without saying that example implementations of the present disclosure can also be operated with other signal sources which can be combined with chopper amplifiers. A frequency f s  of the switching phases of the spinning current mode of the Hall sensor  130  preferably corresponds to the chopper frequency f chop  of the modulator circuits  412  and demodulator circuits  414 . 
     A differential output of the demodulator circuit  414  coupled to the measurement amplifier  120  can be fed for example to a transconductance amplifier  418  connected downstream, the output of which is in turn fed to an inverting input of an operational amplifier  420  connected downstream. A noninverting input of the operational amplifier  420  is coupled to the external supply signal  140  via a low-pass filter  422  and the voltage divider  150 . An overall amplifier ratiometric measurement signal  424  can be tapped off at the output of the operational amplifier  420 . 
     A further example implementation of the sensor circuit  100  is shown in  FIG.  5   . The integrated sensor circuit  500  in  FIG.  5    differs from the integrated sensor circuit  400  in  FIG.  4    in that an up/down counter or digital integrator  510  and a DAC  512  are coupled to the output of the comparator  408  in order to supply a control signal for the controlled current sources  310 ,  312 . 
       FIG.  6    shows an implementation of an integrated ratiometric sensor circuit  600  in which, in the manner similar to  FIGS.  4  and  5   , the amplifiers  118  and  120  are likewise configured as chopper amplifiers, that is to say each have at least one modulator circuit  412  at the input and at least one demodulator circuit  414  at the output of the respective amplifier. In addition, in accordance with the example implementation in  FIG.  6   , the two (identical) amplifiers  118 ,  120  have folded cascodes in each case. Each amplifier  118  and  120  is a folded cascode having a MOS differential stage  302  and a cascode pair  602  (as designated by the dotted line) at the output of the MOS differential stage  302 . 
     With the chopper frequency f chop , inverting and noninverting components of the ratiometric setpoint signal  116  and of an output signal of the folded cascode (amplifier)  118  are alternately fed to a differential input of a transconductance amplifier  608  in order to output the actuating or control signal  114  for the controlled current sources  310 ,  312 . The control current sources  310  here control source bias currents of the respective MOS differential stages  302  and source bias currents of the respective cascode pair  602 . The controlled current sources  312  here control drain bias currents of the respective cascode pair  602 . In the case of the switch position shown in  FIG.  6    at the transconductance amplifier  608 , the lower line (signal component) of the ratiometric setpoint signal  116  is applied to the inverting input of the transconductance amplifier  608  while the left line (signal component) of the output signal of the folded cascode (amplifier)  118  is applied to the noninverting input of the transconductance amplifier  608 . In the next chopper switching phase, the upper line (signal component) of the ratiometric setpoint signal  116  is applied to the noninverting input of the transconductance amplifier  608 , while the right line (signal component) of the output signal of the folded cascode (amplifier)  118  is applied to the inverting input of the transconductance amplifier  608 . The source and drain bias currents adjusted by way of the control loop  110  are distributed via current mirrors  604  in order to set the same gain in each case for both amplifiers (folded cascodes)  118  and  120 . 
     An optional feedback loop  610  is arranged between an output of the main amplifier (folded cascode)  120  and a transconductance amplifier  418  connected downstream, and can feed back the chopper ripple (instead of a largely capacitively coupled chopper demodulator). To put it more precisely, the SC feedback loop  610  can be coupled between an input of the cascode pair  602  (or an output of the MOS differential stage  302 ) and an output of the cascode pair  602  of the main amplifier  120 . In the SC feedback loop  610 , one or more capacitances can be connected between the respective signal line and ground with a switchover frequency f s  for each of the differential signal paths. 
     The circuit arrangement in  FIG.  6    makes it possible to attain at the output  424  a larger dynamic range (even higher gain possible) and more latitude for common-mode levels. The chopping is effected among the cascodes  602 , which is generally faster. Large output resistances with their parasitic capacitances are no longer situated in the chopper loop, but rather still in the gain loop. 
     In accordance with example implementations, the control loop  110  compares a fixed input voltage  112  (e.g. derived from a bandgap circuit or the internal supply of the sensor) with a divided external supply voltage VDDext (which is ratiometric with respect to VDDext)  116  and thus controls the gain of the sensor amplifier. The gain of the replica amplifier  118  is controlled by the comparison of the fixed input signal  112  with the divided VDDext voltage  116  and adjusts the gain of the replica amplifier  118  and of the sensor amplifier  120 . As a result, technology-dependent fluctuations and temperature fluctuations of imprecise amplifiers can be compensated for. Furthermore, the dynamic effects of chopping (gain reduction effects) can be compensated for. 
     The aspects and features which have been described in association with a specific one of the examples above can also be combined with one or more of the further examples in order to replace an identical or similar feature of this further example or in order additionally to introduce the feature into the further example. 
     It furthermore goes without saying that the disclosure of a plurality of steps, processes, operations or functions disclosed in the description or the claims should not be interpreted as being mandatorily in the order described, unless this is explicitly indicated or absolutely necessary for technical reasons in an individual case. Therefore, the preceding description does not limit the implementation of a plurality of steps or functions to a specific order. Furthermore, in further examples, an individual step, an individual function, an individual process or an individual operation can include a plurality of partial steps, partial functions, partial processes or partial operations and/or be subdivided into them. 
     Where some aspects in the preceding sections have been described in association with a device or a system, these aspects should also be understood as a description of the corresponding method. In this case, for example, a block, a device or a functional aspect of the device or of the system can correspond to a feature, for instance a method step, of the corresponding method. Analogously thereto, aspects described in association with a method should also be understood as a description of a corresponding block, a corresponding element, a property or a functional feature of a corresponding device or a corresponding system. 
     The claims that follow are hereby incorporated in the detailed description, where each claim can be representative of a separate example by itself. Furthermore, it should be taken into consideration that—although a dependent claim refers in the claims to a specific combination with one or more other claims—other examples can also encompass a combination of the dependent claim with the subject of any other dependent or independent claim. Such combinations are hereby explicitly proposed, provided that in an individual case no indication is given that a specific combination is not intended. Furthermore, features of a claim are also intended to be included for any other independent claim, even if this claim is not directly defined as being dependent on this other independent claim.