Patent Publication Number: US-11387816-B2

Title: Power controller

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     The present application is based on PCT filing PCT/JP2019/003283, filed Jan. 31, 2019, the entire contents of which are incorporated herein by reference. 
     TECHNICAL FIELD 
     The present invention relates to a power controller. 
     BACKGROUND ART 
     A power controller is used to control an amount of electric power supplied to a load. For example, a satellite uses a power controller as a bus power supply to supply a stabilized voltage of about 50 V or 100 V to its device. For example, a power controller for a satellite supplies a load with electric power generated by multiple photovoltaic arrays during daylight hours and also short-circuits (hereinbelow, also referred to as shunts) an output from a particular photovoltaic array, thereby suppressing an increase in the voltage of the power bus. 
     In such a power controller, the number of stages of photovoltaic arrays depends on the magnitude of the required electric power. Conventional photovoltaic arrays are composed of about 10 stages to about 40 stages, and each photovoltaic array is connected in parallel with a switch element for shunting. In other words, as many switch elements for shunting as the steps of a photovoltaic array are provided. Note that the number of stages of the photovoltaic arrays and switch elements is referred to as a shunt stage count. 
     Such a power controller of shunt system drives switch elements formed of field-effect transistors connected in parallel with the respective photovoltaic arrays to turn on or off, thereby switching between shunting and opening. An operation of switching between turning on and off of the switch element (hereinbelow, referred to as switching) involves heat generation due to a switching loss. In design of a power controller, thermal design is made such that the power controller can withstand a condition of maximum heat generation by switch elements. Accordingly, the power controller has a larger size as a maximum amount of heat generation assumed in individual switch elements is higher. 
     Thus, a system is devised that can make heat generation of switch elements uniform to reduce a maximum amount of heat generation of individual switch elements for a smaller size and a lighter weight of a power controller (for example, see PTL 1). In this system, the power controller determines a ratio between supplying and shunting of electric power (shunt rate) for each control cycle and based on an amount of excess or deficiency of electric power supplied, and calculates one-times of the switch elements based on this shunt rate. The switch elements are sequentially driven in response to the respective timing signals equally assigned. This makes switching counts of all switch elements uniform, and the switching count of an individual switch element is reduced down to a fraction of the shunt stage count, compared to the switching performed in the entire power controller. This can make the heat generated in the switch elements uniform to reduce a maximum amount of heat generation. The condition of thermal design can thus be relaxed, leading to a smaller size and a lighter weight of a power controller. 
     CITATION LIST 
     Patent Literature 
     PTL 1: Japanese Patent Laying-Open No. 2014-71554 
     SUMMARY OF INVENTION 
     Technical Problem 
     The power controller disclosed in PTL 1 distributes switching performed in the entire power controller to all the switch elements. The switching count of an individual switch element is reduced to a fraction of the shunt stage count, compared to the number of switching in the power controller. On the other hand, in order to achieve this operation, the power controller determines the on-times of the switch elements sequentially one by one and drives the switch elements, for each of the control cycles determined by the equally assigned timing signals. Consequently, a time equivalent to the shunt stage count for each control cycle is necessary for driving all the switch elements, resulting in poor responsiveness. 
     As the shunt rate needs to be changed to maintain the bus voltage, for example, as the electric power consumed by a load varies, a time equivalent to the shunt stage count for each control cycle is necessary for changing the shunt rates of all the switch elements. This may cause a control delay dependent on the shunt stage count, and in some cases, fail to maintain the bus voltage, leading to destruction of the apparatus. 
     In order to keep fluctuations of the bus voltage due to such a control delay within an acceptable range, a measure needs to be taken against, for example, an increased capacity of a capacitor (bus capacitor) connected in parallel with a power bus. This may hinder miniaturization of the power controller. 
     The control delay can be reduced by a reduced control cycle, thereby improving responsiveness. This, however, increases the switching count and also increases an amount of heat generated by an individual switch element in proportion to the switching count. This counters an effect of reducing a maximum amount of heat generation, obtained by making heat generation uniform, which may hinder miniaturization of the power controller. 
     The present invention has been made to solve the above problem and has an object to provide a power controller with a smaller size and a lighter weight that is able to reduce a maximum amount of heat generated in an individual switch element and resolve a control delay dependent on the shunt stage count. 
     Solution to Problem 
     The present invention relates to a power controller connected to a plurality of power supplies and a load. The power controller includes: a plurality of switch elements provided corresponding one-to-one to the plurality of power supplies, each of the plurality of switch elements switching on or off to switch between supplying electric power from a corresponding one of the plurality of power supplies to the load and stopping the supply; an operation processing unit to compute an amount of operation for adjusting the electric power supplied to the load; and a signal generator to compute, based on the amount of operation, a number of switch elements to be turned on among the plurality of switch elements and a duty ratio set to the number of switch elements to be turned on and generate, based on the determined number of switch elements and the determined duty ratio, a signal for driving at least one of the plurality of switch elements. The signal generator includes a correction value operation unit to obtain a correction value based on a difference of an on-pulse width between a shunt current flowing through a corresponding one of the plurality of switch elements and a shunt drive signal for driving the corresponding one of the plurality of switch elements, and a corrector to correct, based on the correction value, an amount of operation output from the operation processing unit. 
     Advantageous Effects of Invention 
     The present invention can reduce a maximum amount of heat generated in an individual switch element and resolve a control delay dependent on a shunt stage count, leading to a smaller size and a lighter weight of a power controller. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a circuit diagram of a power controller according to Embodiment 1. 
         FIG. 2  is a block diagram showing an internal configuration of a signal generator  4  of the power controller according to Embodiment 1. 
         FIG. 3  shows an example internal configuration of a modulator  10  of Embodiment 1. 
         FIG. 4  is a timing chart showing time changes of a shunt amount NDUTY_C, a switch-on count MLPWM, and states of switch elements S 1  to S 4 . 
         FIG. 5  is a timing chart showing time changes of shunt amount NDUTY_C, an integral part Int, a carrier wave cw 1 , a fractional part PWM signal FPWM, and switch-on count MLPWM. 
         FIG. 6  is a timing chart showing time changes of an offset value OFFSET and the states of switch elements S 1  to S 4 . 
         FIG. 7  shows waveforms of a shunt drive signal and a shunt current. 
         FIG. 8  is a block diagram showing example configurations of a correction value operation unit  13  and a corrector  9 . 
         FIG. 9  is a timing chart showing signals transmitted in correction value operation unit  13  and corrector  9 . 
         FIG. 10  is a timing chart showing signals transmitted in correction value operation unit  13  and corrector  9 . 
         FIG. 11  shows a simulation result of a waveform of a bus voltage Vbus when pulse width correction control is not performed. 
         FIG. 12  shows a simulation result of a waveform of bus voltage Vbus when pulse width correction control is performed. 
         FIG. 13  shows examples of a modulator  10  and a driving determiner  410  according to a variation of Embodiment 1. 
         FIG. 14  is a timing chart showing waveforms of signals transmitted in modulator  10  and driving determiner  410  of  FIG. 13 . 
         FIG. 15  is a timing chart showing waveforms of signals transmitted in modulator  10  and driving determiner  410  of  FIG. 13 . 
         FIG. 16  is a block diagram showing an internal configuration of a signal generator  4  of a power controller  1  according to Embodiment 2. 
         FIG. 17  is a timing chart showing time changes of a shunt command value SP, outputs xl to xn of JK flip-flops FFd 1  to FFdn, and a switch-on count MLPWM. 
         FIG. 18  is a block diagram showing a part of an internal configuration of a signal generator  4  of a power controller  1  according to a variation of Embodiment 2. 
         FIG. 19  shows a power controller of the variation 
         FIG. 20  shows a configuration of a power controller when functions of an operation processing unit  3 A and signal generator  4  are implemented using software. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Embodiments will be described below with reference to the drawings. 
     Embodiment 1 
       FIG. 1  is a circuit diagram showing a power controller according to Embodiment 1. 
     A power controller  1  is connected to a plurality of direct-current (DC) power supplies I 1  to In (n is an integer not less than 3), which supply electric power, and a load  5 . Power controller  1  controls an amount of electric power supplied from DC power supplies I 1  to In to load  5 . 
     DC power supplies I 1  to In supply electric power to power controller  1 . DC power supplies I 1  to In are formed of, for example, photovoltaic arrays, which may be other power supplies that supply electric power. Although it is assumed in the present embodiment that DC power supplies I 1  to In are mounted in a satellite, the present invention is not limited thereto. DC power supplies I 1  to In may be mounted in other spacecraft such as artificial planets or space stations, as well as apparatuses on the ground, on the sea, or in the air. 
     Power controller  1  includes a plurality of switch elements S 1  to Sn formed of, for example, field-effect transistor (FETs), backflow prevention elements D 1  to Dn formed of diodes, a power bus  2 , an operation processing unit  3 A, a signal generator  4 , and a bus capacitor Cbus. 
     Switch elements S 1  to Sn are provided corresponding one-to-one to DC power supplies I 1  to In. Switch elements Si (i=1 to n) switch on or off to switch between supplying electric power from their corresponding DC power supplies Ii (i=1 to n) to load  5  and stopping the supply. Switch elements Si to Sn are connected in parallel with DC power supplies I 1  to In, respectively. Thus, switch elements Si (i=1 to n) can be turned on to shunt their corresponding DC power supplies Ii (i=1 to n), thus stopping supply of electric power to load  5 . Switch elements Si (i=1 to n) can be turned off to stop the supply of electric power from their corresponding DC power supplies Ii (i=1 to n) to load  5 . 
     Although switch elements Si to Sn are each formed of, for example, a switch element including a field-effect transistor (FET), the present invention is not limited thereto. Switch elements S 1  to Sn may be formed of other types of switch elements. Although switch elements S 1  to Sn are connected in parallel with DC power supplies I 1  to In, the present invention is not limited to such a configuration. It suffices that switch elements Si (i=1 to n) can be turned on or off to supply electric power from their corresponding DC power supplies Ii (i=1 to n) to load  5  or stop the supply. 
     Backflow prevention elements D 1  to Dn are provided corresponding one-to-one to DC power supplies I 1  to In and are connected in series with DC power supplies I 1  to In, respectively. Backflow prevention elements D 1  to Dn are each formed of a diode. Backflow prevention elements D 1  to Dn prevent a backflow of current into DC power supplies I 1  to In. Backflow prevention elements D 1  to Dn are examples of elements that prevent a backflow of current into DC power supplies I 1  to In and may be replaced by other elements having similar functions. 
     DC power supplies Ii (i=1 to n) are connected in parallel between drain terminals and source terminals of their corresponding switch elements Si (i=1 to n). DC power supplies Ii (i=1 to n) are connected to power bus  2  via their corresponding backflow prevention elements Di (i=1 to n). Connection nodes NDi (i=1 to n) between positive electrodes of DC power supplies Ii (i=1 to n) and drain terminals of switch elements Si (i=1 to n) are connected to anode terminals of backflow prevention elements Di (i=1 to n). Cathode terminals of backflow prevention elements D 1  to Dn are connected to power bus  2 . Bus capacitor Cbus and load  5  are connected in parallel to power bus  2 . 
     Operation processing unit  3 A generates a shunt command value SP in accordance with a voltage of power bus  2 . Shunt command value SP is a command value of an amount of operation of shunting performed by the entire power controller  1 , that is, a command value of an amount of operation for adjusting electric power supplied to load  5 . 
     Operation processing unit  3 A generates shunt command value SP based on a difference value between bus voltage Vbus of power bus  2  and a predetermined target control value Vref. The generated shunt command value SP is sent to signal generator  4 . 
     Signal generator  4  is connected to switch elements S 1  to Sn and operation processing unit  3 A. Signal generator  4  determines the number of switch elements to be turned on among switch elements Si to Sn and a duty ratio to be set for the number of switch elements to be turned on, for each control cycle and based on shunt command value SP input from operation processing unit  3 A. The duty ratio is a ratio of an on-time to a control cycle of a switch element to be turned on. 
     Signal generator  4  has output terminals connected to gate terminals of switch elements Si to Sn. Switch elements Si to Sn are driven to turn on or off in response to a drive signal output from signal generator  4 . 
     Load  5  is, for example, a device mounted in a satellite and is connected to power controller  1 . Load  5  may be, for example, an electrical storage device, such as a battery, and may be connected via a charge-discharge controller. 
     Power controller  1  described in Embodiment 1 supplies electric power from DC power supplies I 1  to In via power bus  2  to load  5 . A voltage supplied to load  5  is maintained by bus capacitor Cbus. Power controller  1  supplies electric power generated from DC power supplies I 1  to In to load  5  during daylight hours, while short-circuiting (shunting) a surplus of the generated electric power at appropriate time intervals and at any appropriate ratio, thereby controlling bus voltage Vbus to suppress an increase in bus voltage Vbus. 
     An operation of power controller  1  in the present embodiment will now be described. In Embodiment 1, operation processing unit  3 A outputs shunt command value SP in order to decrease a difference between bus voltage Vbus and the predetermined target control value Vref, thereby controlling bus voltage Vbus to attain to a certain voltage. Shunt command value SP is also an amount of operation for adjusting an amount of electric power supplied to load  5 . 
     Although an operation of operation processing unit  3 A will be described below by taking, as an example, a system in which a difference between bus voltage Vbus and target control value Vref is a deviation Error and PID (proportional-integral-derivative) control is performed, the present invention is not limited to such a system. Although target control value Vref is a rated value of bus voltage Vbus, which is generally 50 V to 100 V, the present invention is not limited to this value. 
     Operation processing unit  3 A includes a subtractor  31  and a PID controller  32 . 
     Subtractor  31  calculates deviation Error indicating a difference between bus voltage Vbus of power bus  2 , detected by a voltage detector (not shown), and the predetermined target control value Vref. Deviation Error is expressed by Equation (1).
 
Error= V bus− V ref  (1)
 
     PID controller  32  performs proportional, integral, and derivative operations on deviation Error, and outputs a signal obtained by adding the results thereof as shunt command value SP. Specifically, shunt command value SP is expressed by Equation (2) using constants KP, KI, and KD. 
     
       
         
           
             
               
                 
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     Appropriate numeric values not less than 0 are selected for constants KP, KI, and KD in Equation (2) so as to allow for target control with a constant bus voltage in accordance with a circuit constant. Shunt command value SP is a numeric value indicating, in each control cycle, an average number of switch elements to be shunted. Shunt command value SP is any appropriate value of 0 to n in the power controller described in Embodiment 1. It means that the entire generated power is supplied to load  5  when shunt command value SP is 0, and the entire generated power is shunted when shunt command value SP is n to stop supply of electric power to load  5 . 
     Shunt command value SP computed by operation processing unit  3 A is transmitted to signal generator  4 . Signal generator  4  determines the number of switch elements to be turned on (hereinbelow, referred to as a switch-on count) and a ratio of an on-time to a control cycle of switch elements to be turned on (hereinbelow, referred to as a duty ratio) in order to achieve shunting specified by the input shunt command value SP. 
     Signal generator  4  drives at least one of switch elements S 1  to Sn based on the switch-on count. Signal generator  4  exchanges a switch element to be driven as the switch-on count decreases. In other words, a switch element different from a switch element which has been driven last time is driven. 
       FIG. 2  is a block diagram showing an internal configuration of signal generator  4  of the power controller according to Embodiment 1. 
     Signal generator  4  includes a timing signal generator  8 , a sampling unit  7 , a corrector  9 , a modulator  10 , an offset generator  11 , an overvoltage detector  12 , a correction value operation unit  13 , and a driving determiner  410 . 
     Timing signal generator  8  outputs a timing signal Tsmpl. 
     Sampling unit  7  samples shunt command value SP of  FIG. 1  in cycles of timing signal Tsmpl, thereby generating a shunt amount NDUTY, which is an amount of operation for adjusting electric power supplied to load  5 . 
     Corrector  9  corrects shunt amount NDUTY and outputs a corrected shunt amount NDUTY_C. 
     Modulator  10  outputs a switch-on count MLPWM based on the corrected shunt amount NDUTY_C. When the corrected shunt amount NDUTY_C is an integral value, modulator  10  outputs the integral value as switch-on count MLPWM. When the corrected shunt amount NDUTY_C is not an integral value, modulator  10  outputs the sum of an integral part Int and FPWM, which will be described below, as switch-on count MLPWM. 
       FIG. 3  shows an example internal configuration of modulator  10  of Embodiment 1. 
     Modulator  10  includes a carrier wave generator  401 , a divider  406 , a comparator  403 , and an adder  405 . 
     Carrier wave generator  401  outputs a carrier wave cw 1 . 
     Divider  406  divides the corrected shunt amount NDUTY_C output from corrector  9  into integral part Int and a fractional part Frac. Since the effective range of shunt amount NDUTY_C is 0 to n, integral part Int is an integral value of 0 to n, and fractional part Frac is a fractional value not less than 0 and less than 1. 
     Comparator  403  compares fractional part Frac with carrier wave cw 1 , thereby generating a fractional part PWM (Pulse Width Modulation) signal FPWM. Fractional part PWM signal FPWM assumes a value of 0 or 1 at any appropriate duty ratio. An average of fractional part PWM signal FPWM is fractional part Frac. 
     Adder  405  adds integral part Int and fractional part PWM signal FPWM (0 or 1) together, thereby calculating switch-on count MLPWM. The calculated switch-on count MLPWM is sent to driving determiner  410  of  FIG. 2  and is also sent to offset generator  11 . 
     Referring again to  FIG. 2 , offset generator  11  converts switch-on count MLPWM into an offset value OFFSET. Offset value OFFSET is sent to driving determiner  410 . 
     Upon detection of a decrease in switch-on count MLPWM, offset generator  11  performs modulo addition of an amount of the decrease and offset value OFFSET modulo n, thereby updating offset value OFFSET. Offset value OFFSET is a value indicating a starting point of switch elements to be driven and assumes an integral value of 0 to (n−1). Switch element Si is a starting point when offset value OFFSET is 0, and switch element S 2  is a starting point when offset value OFFSET is 1. 
     Since the determined switch-on count MLPWM and offset value OFFSET fix the number of switch elements to be turned on among switch elements Si to Sn and a starting point of the switch elements to be turned, the driving state (on/off) of switch elements Si to Sn can be determined uniquely. 
     Driving determiner  410  includes determiners  411 - 1  to  411 - n.    
     Determiner  411 - i  (i=1 to n) determines whether switch element Si (i=1 to n) belongs to the range where the switch element Si is to be turned on, and determines whether to drive switch element Si. In an example determination method, switch element Si with a shunt number i can be driven when any of Inequalities (3) and (4) composed of shun number i, shunt stage count n, switch-on count MLPWM, and offset value OFFSET is satisfied.
 
OFFSET≤ i &lt;(OFFSET+MLPWM)  (3)
 
OFFSET≤( n+i )&lt;(OFFSET+MLPWM)  (4)
 
       FIG. 4  is a timing chart showing time changes of shunt amount NDUTY_C, switch-on count MLPWM, and the states of switch elements Si to S 4 .  FIG. 5  is a timing chart showing time changes of shunt amount NDUTY_C, integral part Int, carrier wave cw 1 , fractional part PWM signal FPWM, and switch-on count MLPWM.  FIG. 6  is a timing chart showing time changes of offset value OFFSET and the states of switch elements S 1  to S 4 . 
     In  FIGS. 4 to 6 , the horizontal axis represents a lapse of time, and the vertical line represents the state of each signal. Although the case where shunt stage count n is 4 is illustrated here by way of example for ease of explanation, the present invention is not limited to this value. 
     I 1  lustrated here is an example operation when 2.4 and 1.6 are provided as the corrected shunt amount NDUTY_C. 
     In the first and second control cycles, 2.4 is input as the corrected shunt amount NDUTY_C. 
     Divider  406  divides the corrected shunt amount NDUTY_C(=2.4) output from corrector  9  into integral part Int (=2) and fractional part Frac (=0.4). Comparator  403  compares fractional part Frac (=0.4) with carrier wave cw 1  generated in carrier wave generator  401 , thereby generating fractional part PWM signal FPWM. Adder  405  adds integral part Int (=2) and fractional part PWM signal FPWM (0 or 1, average=0.4) together, thereby calculating switch-on count MLPWM (2 or 3). 
     Thus, switch-on count MLPWM=3 in 40% of control cycle T, and switch-on count MLPWM=2 in 60% of control cycle T. An average of switch-on counts MLPWM in the first and second control cycles is 2.4. 
     Switch-on count MLPWM is 3 at an initial timing of the first control cycle. As a result, three switch elements S 1 , S 2 , and S 3  are turned on starting from switch element S 1 . 
     Switch-on count MLPWM decreases to 2 at a subsequent timing of the first control cycle. Offset generator  11  performs modulo addition of an amount of the decrease (=1) of switch-on count MLPWM and offset value OFFSET (0) modulo 4, thereby updating offset value OFFSET to “1”. Thus, two switch elements are turned on starting from switch element S 2  while rotating switch elements to be driven. Specifically, switch elements S 2  and S 3  are turned on. 
     At an initial timing of the second control cycle, switch-on count MLPWM increases to 3, and accordingly, three switch elements are turned on while keeping switch element S 2  as the starting point, without rotation. Specifically, switch elements S 2 , S 3 , and S 4  are turned on. 
     Subsequently, switch-on count MLPWM decreases to 2 at a subsequent timing of the second control cycle. Offset generator  11  performs modulo addition of an amount of the decrease (=1) of switch-on count MLPWM and offset value OFFSET (1) modulo 4, thereby updating offset value OFFSET to “2”. Thus, two switch elements are turned on starting from switch element S 3 . Specifically, switch elements S 3  and S 4  are turned on. 
     In the third and fourth control cycles, 1.6 is input as a shunt command value. Divider  406  divides the corrected shunt amount NDUTY_C(=1.6) output from corrector  9  into integral part Int (=1) and fractional part Frac (=0.6). Comparator  403  compares fractional part Frac (=0.6) with carrier wave cw 1  generated in carrier wave generator  401 , thereby generating fractional part PWM signal FPWM. Adder  405  adds integral part Int (=1) and fractional part PWM signal FPWM (0 or 1, average=0.6) together, thereby calculating switch-on count MLPWM (1 or 2). 
     Thus, switch-on count MLPWM=2 in 60% of control cycle T, and switch-on count MLPWM=1 in 40% of control cycle T. An average of switch-on counts MLPWM in the third and fourth control cycles is 1.4. 
     The state where switch-on count MLPWM is 2 continues at an initial timing of the third control cycle. Thus, a state in which switch elements S 3  and S 4  are turned on and switch elements S 1  and S 2  are turned off continues. 
     Switch-on count MLPWM decreases to 1 at a subsequent timing of the third control cycle. Offset generator  11  performs modulo addition of an amount of the decrease (=1) of switch-on count MLPWM and offset value OFFSET (2) modulo 4, thereby updating offset value OFFSET to “3”. Thus, one switch element is turned on starting from switch element S while rotating switch elements to be driven. Specifically, switch element S 4  is turned on. 
     At an initial timing of the fourth control cycle, switch-on count MLPWM increases to 2, and accordingly, two switch elements are turned on while keeping switch element S 4  as the starting point, without rotation. Specifically, switch elements S 4  and S 1  are turned on. 
     Switch-on count MLPWM decreases to 1 at a subsequent timing of the fourth control cycle. Offset generator  11  performs modulo addition of an amount of the decrease (=1) of switch-on count MLPWM and offset value OFFSET (3) modulo 4, thereby updating offset value OFFSET to “0”. Thus, one switch element is turned on starting from switch element S 0  while rotating a switch element to be driven. Specifically, switch element S 1  is turned on. 
     As a switch element which is a starting point to be turned on for each control cycle changes, the switching operation can be distributed to all of switch elements S 1  to S 4 , as described above. Accordingly, a timing of switching one switch element can be reduced to one time in four control cycles  4 T. 
     As a result, when considering power controller  1  in which the shunt stage count is n as a whole, the switching operation of each of switch elements S 1  to Sn is performed for every n control cycles while performing switching in the same cycle as the control cycle. Since a switching count of each of switch elements S 1  to Sn is reduced to 1/n of the switching count of the entire power controller  1 , a heat generation condition required for switch elements S 1  to Sn is relaxed. The switch element is driven by determining the states of all the switch elements for each control cycle, which causes no control delay dependent on shunt stage count n. 
     In the power controller described in Embodiment 1, a switch element which is a starting point is rotated as the switch-on count decreases as described above, and thus, a switch element with the longest on-time among a plurality of turned-on switch elements is turned off. Similarly, switch elements for an amount of switch-on count are turned on from a switch element which is a starting point as the switch-on count increases, and thus, a switch element with the longest off-time among a plurality of turned-off switch elements is turned on. The case where the switch-on count decreases by one is described here, which shows a configuration in which a switch element serving as a starting point is rotated by one. Alternatively, when the switch-on count decreases by two or more, a rotation may be performed correspondingly. 
     In  FIG. 2 , the cycle of timing signal Tsmpl is set to reduce heat generation through switching of switch elements Si to Sn in a stationary operation. 
     A bus voltage control operation is performed accompanied by PID control in cycles of timing signal Tsmpl, and accordingly, a negative feedback voltage control operation is updated for each certain switching cycle. As a result, bus voltage Vbus cannot be controlled for more rapid load fluctuations than the switching cycle, and accordingly, the bus voltage may transiently fluctuate. Overvoltage detector  12  is provided as a measure taken when bus voltage Vbus fluctuates greatly due to sudden fluctuations of load  5 . 
     When a fluctuation range of bus voltage Vbus, which is a voltage supplied to load  5 , exceeds a preset fluctuation range due to fluctuations of load  5 , overvoltage detector  12  outputs overvoltage detection signal Trn to timing signal generator  8 . Upon receipt of overvoltage detection signal Trn, timing signal generator  8  enters a transient response state, thus reducing the cycle of timing signal Tsmpl. This leads to a more rapid bus voltage control response, thus preventing transient occurrence of large fluctuations of bus voltage. 
     After fluctuations of bus voltage Vbus converge, timing signal generator  8  waits for a time that elapses before heat generation of switch elements Si to Sn is less affected. Timing signal generator  8  then returns the cycle of timing signal Tsmpl to the cycle in the original stationary operation. This reduces the cycle of timing signal Tsmpl by a period of the transient response state, thus preventing heat generation of switch elements Si to Sn from being affected. 
     In the above operation, however, at a timing at which timing signal Tsmpl with a reduced cycle returns to an original cycle after fluctuations of bus voltage Vbus converge, bus voltage Vbus may fluctuate again. 
       FIG. 7  shows waveforms of a shunt drive signal and a shunt current. 
     As shown in  FIG. 7 , a shunt current actually flowing through switch elements Si to Sn may delay with respect to rising and falling of shunt drive signals of switch elements Si to Sn. Specifically, a delay time ON_Delay in turn-on may be different from a delay time OFF_Delay in turn-off. In such a case, ON_Duty, which is a ratio of an on-time to a switching cycle, and effective ON_Duty, which is a ratio of an on-time to a switching cycle of an effective shunt current, differ dependent on the cycle of timing signal Tsmpl. Since effective ON_Duty changes at a timing of switching of the cycle of timing signal Tsmpl, bus voltage Vbus fluctuates. Such fluctuations return through a feedback control operation of bus voltage Vbus after a lapse of a response time of a control system. Correction value operation unit  13  is provided as a measure taken against this phenomenon. 
     Correction value operation unit  13  performs a correction operation on a difference between effective ON_Duty, which changes dependent on the cycle of timing signal Tsmpl, and ON_Duty of a shunt drive signal, using a value regarded as correction value ΔNDUTY in advance regardless of a bus voltage control response operation. This can prevent a phenomenon in which bus voltage Vbus fluctuates again at a timing at which timing signal Tsmpl returns to an original cycle after fluctuations of bus voltage Vbus converge. 
     When a delay time during ON of a shunt current is ON_Delay and a delay time during OFF is OFF_Delay with respect to a shunt drive signal, a difference ΔP W  of an on-pulse width between a shunt current and a shunt drive signal is expressed by Equation (5). The difference ΔP W  is a difference between an on-pulse width of a shunt current and an on-pulse width of a shunt drive signal.
 
Δ P   W =ON_Delay−OFF_Delay  (5)
 
     Correction value ΔNDUTY, which is an amount of change of shunt amount NDUTY caused by a difference ΔP W  of the on-pulse width, is expressed by Equation (6) where the cycle of timing signal Tsmpl is T.
 
Δ N DUTY= n×ΔP   W   /T   (6)
 
     Correction value operation unit  13  calculates correction value ΔNDUTY in accordance with Equations (5) and (6). 
     A difference ΔP W  of on-pulse width expressed by Equation (5) is a value dependent on the characteristics of switch elements Si to Sn and a design condition of a shunt driving unit. A value of difference ΔP W  of on-pulse width thus may be a numeric value determined in design of an apparatus or a numeric value measured in release testing. It is conceivable that difference ΔP W  of on-pulse width may fluctuate though it fluctuates by a small amount. Such fluctuations result from temperature fluctuations, a change over the years, or the like, resulting in gradual fluctuations. In the case of gradual fluctuations, correction value operation unit  13  periodically measures difference ΔP W  of on-pulse width, and calculates correction value ΔNDUTY for each cycle T of timing signal Tsmpl in accordance with Equation (6). 
     Corrector  9  corrects shunt amount NDUTY to NDUTY_C in accordance with Equation (7).
 
 N DUTY_ C=N DUTY−Δ N DUTY  (7)
 
     This also enables a correction operation on shunt amount NDUTY in synchronization with a switch of cycle T regardless of the bus voltage control response operation. 
     When large fluctuations of difference ΔP W  of on-pulse width are accommodated, for example, operations of Equations (5) and (6) may be performed based on a representative shunt drive signal Gn and a differential detection signal Diff of shunt current Ishn to calculate correction value ΔNDUTY. Shunt drive signal Gn is a signal for driving switch element Sn. Shunt current Ishn is a current flowing through switch element Sn. Differential detection signal Diff is obtained from, for example, simple current detector  14  including a current transformer or the like. 
     Next, specific configurations of corrector  9  and correction value operation unit  13  in  FIG. 2  will be described. 
       FIG. 8  is a block diagram showing example configurations of correction value operation unit  13  and corrector  9 . 
       FIGS. 9 and 10  are timing charts showing signals transmitted in correction value operation unit  13  and corrector  9 . 
     A main part of the operation of obtaining difference ΔP W  of on-pulse width and a corrected shunt amount NDUTY_C will be described with reference to  FIGS. 8 to 10 . 
     A comparator CmpH compares differential detection signal Diff that has a differentiated waveform, detected from shunt current Ishn, with a reference value Ref_H, and outputs a result of the comparison as an on-timing signal Comp_H of shunt current Ishn. 
     Comparator CmpL compares differential detection signal Diff that has a differentiated waveform, detected from shunt current Ishn, with a reference value Ref_L and outputs a result of the comparison as an off-timing signal Comp_L of shunt current Ishn. 
     A JK flip-flop JK includes a J input terminal that receives on-timing signal Comp_H of shunt current Ishn and a K input terminal that receives off-timing signal Comp_L of shunt current Ishn and generates a shunt current synchronization signal Ishunt. 
     Operations described hereinbelow are performed in synchronization with a clock signal Sysclock. 
     A counter CT 1  receives shunt drive signal Gn corresponding to shunt current Ishn and counts a pulse number of clock signal Sysclock during an ON period of shunt drive signal Gn. 
     A flip-flop FF 1  holds an output from counter CT 1  in response to shunt drive signal Gn to output a count value a during the ON period of shunt drive signal Gn. 
     A counter CT 2  receives shunt current synchronization signal Ishunt corresponding to shunt current Ishn and counts a pulse number of clock signal Sysclock during the ON period of shunt current synchronization signal Ishunt. 
     A flip-flop FF 2  holds an output from counter CT 2  in response to shunt current synchronization signal Ishunt to output a count value b during an ON period of shunt current synchronization signal Ishunt. 
     A subtractor Sub 1  subtracts count value b during the ON period of shunt current synchronization signal Ishunt from count value a during the ON period of shunt drive signal Gn and outputs a difference d. 
     Counter CT 1  and counter CT 2  each include a zero-load input L and perform a count operation during the ON periods of shunt drive signal Gn and shunt current synchronization signal Ishunt, accompanied by NAND circuits ND 1  and ND 2 . 
     A NOR circuit NOR outputs a timing signal c indicating a period in which both of shunt drive signal Gn and shunt current synchronization signal Ishunt are OFF when overvoltage detection signal Trn is not output, that is, during the stationary operation. 
     A flip-flop FF 3  holds a difference d between the count values in accordance with timing signal c to generate a difference ΔP W  of the on-pulse width. Specifically, difference ΔP W  of the on-pulse width due to difference d of the count value is generated during a period in which both of shunt drive signal Gn and shunt current synchronization signal Ishunt are OFF in the state where overvoltage detection signal Trn is not output, that is, during the stationary operation. 
     A selector SL selects a coefficient C_trn or C_Norm in accordance with whether it is a time at which overvoltage detection signal Trn is output. Coefficient C_trn is a coefficient corresponding to the time at which overvoltage detection signal Trn is output. Coefficient C_Norm is a coefficient corresponding to a time at which overvoltage detection signal Trn is not output. Coefficient C_trn and coefficient C_Norm are expressed by Equations (8) and (9), respectively, from Equations (6) using a total number n of switch elements Si to Sn.
 
 C _ trn=n /Ttrn  (8)
 
 C _Norm= n /Tnrm  (9)
 
     Ttrn in Equation (8) represents a cycle of timing signal Tsmpl when overvoltage detection signal Trn is output, that is, in the transient response state. Tnrm in Equation (9) represents a cycle of timing signal Tsmpl when overvoltage detection signal Trn is not output, that is, in the stationary operation. The following is satisfied. Specifically, the length of the control cycle in the transient response state is shorter than the length of the control cycle in the stationary operation.
 
Ttrn&lt;Tnrm  (10)
 
     A multiplier Mull multiplies difference ΔP W  of the on-pulse width by an output from selector SL, thereby generating a correction value ΔNDUTY. 
     A subtractor Sub 2  generates a shunt amount NDUTY_C obtained by correction through subtraction of correction value ΔNDUTY from shunt amount NDUTY. 
     Since fluctuations of difference ΔP W  of the on-pulse width are small as described above, a fixed value can be used when such fluctuations can be ignored. In such a case, the product of C_trn and ΔP W  and the product of C_Norm and ΔP W  are calculated in advance and provided, leading to a reduced size of the circuit. 
     Since a coefficient to be maintained can be reduced to one by using the numerical values expressed by Equations (11) and (12) as coefficients C_trn and C_Norm, similar effects can be obtained while reducing the size of the circuit.
 
 C _ trn =( n /Ttrn)−( n /Tnrm)  (11)
 
 C _Norm=0  (12)
 
       FIG. 11  shows a simulation result of a waveform of bus voltage Vbus when pulse width correction control is not performed.  FIG. 12  shows a simulation result of the waveform of bus voltage Vbus when pulse width correction control is performed. 
     The simulation result is obtained by simulating fluctuations of bus voltage Vbus when the state in which the cycle of timing signal Tsmpl is shorter in the transient response state is returned to the state in which the normal cycle of timing signal Tsmpl. The cycle of timing signal Tsmpl is switched in the vicinity of 15% of the horizontal axis. It is found that when pulse width correction control is not performed, bus voltage Vbus decreases due to fluctuations of effective ON_Duty. When pulse width correction control described in Embodiment 1 is performed, effective ON_Duty can be kept constant, thus reducing fluctuations of bus voltage Vbus. 
     In the power controller described in the present embodiment, the power controller that performs control with a constant bus voltage obtains a shunt command value, which is an average of the numbers for shunt, from a deviation between a bus voltage and a target value, and rotates a switch element to be driven, at a timing at which the switch-on count increases or decreases while controlling a switch-on count of switch elements and a duty ratio, as described above. This causes no delay dependent on shunt stage count n while reducing the switching frequency of an individual switch element to 1/n, leading to high responsiveness. Further, the switching cycle is made variable in accordance with the magnitude of load fluctuations, thus improving a response during load fluctuations while reducing heat generation in the stationary state. Additionally, fluctuations of a bus voltage can be reduced by correcting effective ON_Duty generated in switch of the switching cycle. 
     The present embodiment can thus reduce heat generation associated with switching while maintaining the stability of a bus voltage. The present embodiment can thus relax the condition for selecting components and heat dissipation design. Consequently, a power controller can be provided that has a high response, a smaller size, a smaller weight, and an excellent electrical power quality. 
     Variation of Embodiment 1 
     Power controller  1  of a variation of Embodiment 1 is obtained by modifying the internal configurations of modulator  10  and driving determiner  410  in power controller  1  of Embodiment 1. The other configuration of the power controller of the variation of Embodiment 1 is similar to that of  FIG. 1 , which will not be described repeatedly. 
       FIG. 13  shows examples of modulator  10  and driving determiner  410  of the variation of Embodiment 1. The case where shunt stage count n is 4 will be described here for ease of explanation. 
     Modulator  10  includes carrier wave generator  401 , offset superimposing units  402 - 1  to  402 - 3 , comparators  403 - 1  to  403 - 4 , and an adder  405 . 
     Driving determiner  410  includes an offset generator  11  and a drive signal assignment unit  409 . 
       FIGS. 14 and 15  are timing charts showing the waveforms of signals transmitted in modulator  10  and driving determiner  410  of  FIG. 13 . 
     Carrier wave generator  401  generates a carrier wave cw 1  having an amplitude of  1  and sends carrier wave cw 1  to comparator  403 - 1  and offset superimposing unit  402 - 1 . 
     Offset superimposing unit  402 - 1  sends, to offset superimposing unit  402 - 2  and comparator  403 - 2 , a signal cw 2  obtained by adding an offset equal to the amplitude of carrier wave cw 1  to carrier wave cw 1 . 
     Offset superimposing unit  402 - 2  sends, to offset superimposing unit  402 - 3  and comparator  403 - 3 , a signal cw 3  obtained by adding an offset equal to the amplitude of carrier wave cw 1  to signal cw 2 . 
     Offset superimposing unit  402 - 3  sends, to comparator  403 - 4 , a signal cw 4  obtained by adding an offset equal to the amplitude of carrier wave cw 1  to signal cw 3 . 
     Comparator  403 - i  (i=1 to 4) compares the corrected shunt amount NDUTY_C with carrier wave cw 1  and outputs a PWM signal Pi. When shunt amount NDUTY_C is greater than carrier wave cw 1 , PWM signal Pi is “1”. When shunt amount NDUTY_C is not greater than carrier wave cw 1 , PWM signal Pi is “0”. An output from comparator  403  with the number corresponding to the switch-on count MLPWM is “1”. 
     This operation yields a PWM signal obtained by dividing a signal domain by the shunt stage count. 
     Adder  405  adds PWM signals P 1  to P 4 , which are outputs from comparators  403 - 1  to  403 - 4 , together and outputs switch-on count MLPWM. 
     Offset generator  11  assigns offset value OFFSET based on switch-on count MLPWM as in Embodiment 1. 
     PWM signal determined so far and offset value OFFSET fix a drive pattern of switch elements and a starting point thereof, and accordingly, the drive state (on/off) of each switch element can be determined uniquely. 
     Drive signal assignment unit  409  performs a cyclic shift operation on PWM signal Pi output from comparator  403 - i  by offset value OFFSET to drive each switch element to be turned on or off. Although  FIG. 13  shows an example in which two-input, one-output multiplexers are combined for cyclic shift, a multi-input multiplexer may be used for cyclic shift. 
     Embodiment 2 
     Power controller  1  of Embodiment 2 is obtained by modifying the internal configuration of signal generator  4  in power controller  1  of Embodiment 1. The other configuration of the power controller according to Embodiment 2 is similar to that of  FIG. 1 , which will not be described repeatedly. 
       FIG. 16  is a block diagram showing an internal configuration of signal generator  4  of power controller  1  of Embodiment 2. 
     Signal generator  4  includes a comparator  17 , a counter  16 , offset generator  11 , and driving determiner  410 . Offset generator  11  and driving determiner  410  are similar to those of Embodiment 1, which will not be described repeatedly. 
     Comparator  17  compares first to n-th on-level thresholds, which are assigned within a variable range of shunt command value SP, and first to n-th off-level thresholds, which are assigned to have a hysteresis corresponding one-to-one to the first to n-th on-level thresholds, with shunt command value SP. 
     Counter  16  determines switch-on count MLPWM based on a result of the comparison of comparator  17 . 
     Comparator  17  includes n number of on-comparators Hii (i=1 to n) and n number of comparators Loi (i=1 to n). 
     Counter  16  includes n number of JK flip-flops FFdi (i=1 to n) and an adder  15 . The JK flip-flops may be replaced by RS flip-flops. 
     An i-th on-comparator Hii compares shunt command value SP with an i-th on-level threshold ONi. The i-th on-comparator Hii outputs “1” when shunt command value SP is greater than the i-th on-level threshold ONi and outputs “0” when shunt command value SP is smaller than the i-th on-level threshold ONi. The i-th on-comparator Hii sends a result of the comparison to a J input terminal of JK flip-flop FFdi of counter  16 . 
     An i-th off-comparator Loi compares shunt command value SP with an i-th off-level threshold OFFi. The i-th off-comparator Loi outputs “1” when the i-th off-level threshold OFFi is greater than shunt command value SP and outputs “0” when the i-th off-level threshold OFFi is smaller than shunt command value SP. The i-th off-comparator Loi sends a result of the comparison to a K input terminal of JK flip-flop FFdi of counter  16 . 
     An i-th JK flip-flop FFdi includes a J input terminal that receives an output from the i-th on-comparator Hii and a K input terminal that receives an output from the i-th off-comparator Loi. 
     An output xi of the i-th JK flip-flop FFdi is sent to adder  15 . 
     Adder  15  adds outputs xl to xn of JK flip-flops FFd 1  to FFdn together and sends an addition value as switch-on count MLPWM to driving determiner  410  and offset generator  11 . 
       FIG. 17  is a timing chart showing time changes of shunt command value SP, outputs xl to xn of JK flip-flops FFd 1  to FFdn, and switch-on count MLPWM.  FIG. 17  also shows on-level thresholds ON 1  to ONn and off-level thresholds OFF 1  to OFFn. The number of stages n of DC power supplies I 1  to In is 5 for ease of explanation. 
     Herein, shunt command value SP is expressed by Equation (13) as a signal indicating an error from target control value Vref in relation to bus voltage Vbus.
 
 SP =( V bus− V ref)× A   (13)
 
     In Equation (12), A is a value indicating an amplification degree of operation processing unit  3 A and is set as an amplification degree with appropriate frequency characteristics, and A is 1 here for ease of explanation. This is because the relation between on-level thresholds ON 1  to ON 5  and off-level thresholds OFF 1  to OFFS, and shunt command value SP is normalized for description. 
     In the relation between on-level thresholds ON 1  to ON 5  and off-level thresholds OFF 1  to OFFS, and shunt command value SP, for example, switch elements S 1 , S 2  are opened (off state) when shunt command value SP falls below off-level thresholds OFF 1 , OFF 2 . Switch elements S 2  to S 5  are shunted (on state) when shunt command value SP exceeds on-level thresholds ON 2  to ON 5 . 
     In other words, on-level thresholds ON 1  to ONn and off-level thresholds OFF 1  to OFFn are set as thresholds with hystereses that determine on and off of switch elements Si to Sn relative to shunt command value SP. 
     In this operation, the setting range of on-level thresholds ON 1  to ONn or the setting range of off-level thresholds OFF 1  to OFFn is a control voltage fluctuation range ΔVbus of bus voltage Vbus. A difference between on-level threshold ONi (i=1 to n) and off-level threshold OFFi (i=1 to n) is a ripple voltage Vripple of bus voltage Vbus. 
     In this operation, bus voltage Vbus fluctuates dependent on a solar battery generation current Istr per stage, bus capacitor Cbus, and load current I 1 oad. This causes shunt command value SP to increase and decrease between on-level thresholds ON 1  to ONn and off-level thresholds OFF 1  to OFFn. A time change ratio of bus voltage Vbusdv/dt is expressed by Equation (14).
 
dV/dt={Istr×( n −MLPWM)− I load}/Cbus  (14)
 
     The value of switch-on count MLPWM changes every time shunt command value SP crosses on-level thresholds ON 1  to ONn and off-level thresholds OFF 1  to OFFn. 
     Specifically, when shunt command value SP falls below off-level threshold OFFi with xi being ON(=1), xi is OFF (=0). When shunt command value SP exceeds on-level threshold ONi with xi being OFF (=0), xi is ON(=1). This changes switch-on count MLPWM. Thus, the bus voltage is controlled while switching ON/OFF of switch elements Si to Sn in turn. 
     Power controller  1  controls a bus voltage such that bus voltage Vbus falls within the range of a defined control voltage fluctuation range ΔVbus, accompanied by ripple voltage Vripple, while dissipating heat generated through switching of switch elements Si to Sn. 
     In the present embodiment, the above operation causes the switching cycles of switch elements Si to Sn to change depending on the state of load  5 . A minimum cycle Tmin in the state in which the switching cycle continues steadily is expressed by Equation (15).
 
 T min=4 ×C bus× V ripple/Istr  (15)
 
     According to Equation (15), thus, minimum cycle Tmin that can be continued steadily can be set within an allowable range of heat generation of switch elements Si to Sn by selecting a related constant and designing power controller  1 . 
     When load current I 1 oad changes greatly in a transient manner, dv/dt is larger in the positive or negative direction according to Equation (14). Consequently, a switching interval of switch elements Si to Sn is shorter, resulting in a more rapid bus voltage control response to sharp load fluctuations. This can prevent large fluctuations of bus voltage in a transient manner. 
     The above principle stabilizes a bus voltage relative to rapid load fluctuations without increasing heat generation by switching of each switch element during steady operation. 
     As described above, in the power controller according to the present embodiment, the comparator compares as many on-level thresholds and off-level thresholds as the shunt stage count, each of which has a hysteresis relative to a fluctuation range of a bus voltage, with the bus voltage, the flip-flop latches a result of the comparison, and the counter adds up the comparison results, thereby calculating a switch-on count MLPWM. 
     During steady loading, thus, the switching cycle is restricted to an appropriate range, and heat generation by switching of a switch element does not increase. 
     During transient load fluctuations, switching intervals are shorter, and a control response can be made also to rapid load fluctuations, thus preventing transient occurrence of large bus voltage fluctuations. Consequently, a power controller with an excellent power quality can be provided without hindering a power controller from having a smaller size and a lighter weight. 
     Variation of Embodiment 2 
     The present variation relates to a case where signal generator  4  is configured by an integrated circuit or software. 
       FIG. 18  is a block diagram showing a part of an internal configuration of signal generator  4  of power controller  1  according to the variation of Embodiment 2. 
     Signal generator  4  includes an adder  551 , a subtractor  552 , a multiplier  553 , a multiplier  555 , an integer formatting unit  554 , an integer formatting unit  556 , a comparator  557 , a comparator  558 , and a counter  559 . 
     Adder  551  adds shunt command value SP and Vrippe/2 together. 
     Subtractor  552  subtracts Vrippe/2 from shunt command value SP. 
     Multiplier  553  multiplies an output from adder  551  by n/ΔVbus. 
     Multiplier  555  multiples an output from subtractor  552  by n/ΔVbus. 
     Integer formatting unit  554  outputs an integral part of an output from multiplier  553 . 
     Integer formatting unit  556  outputs an integral part of an output from multiplier  555 . 
     Comparator  557  compares an output from integer formatting unit  554  with switch-on count MLPWM. Comparator  557  outputs “1” when the output from integer formatting unit  554  is greater than switch-on count MLPWM. Comparator  557  outputs “0” when the output from integer formatting unit  554  is smaller than switch-on count MLPWM. 
     Comparator  558  compares an output from integer formatting unit  556  with switch-on count MLPWM. Comparator  558  outputs “0” when the output from integer formatting unit  556  is greater than switch-on count MLPWM. Comparator  558  outputs “1” when the output from integer formatting unit  554  is smaller than switch-on count MLPWM. 
     Counter  559  increments a count value by an output from comparator  557  and decrements the count value by an output from comparator  558 . Counter  559  outputs switch-on count MLPWM as the count value. 
     The present variation can replace the above operation in which thresholds and comparators required for the shunt stage count with an operation in which thresholds and comparators are not dependent on the stage count, leading to a simplified operation. 
     Variation 
     The present invention is not limited to the above embodiments and includes, for example, the following variations. 
     (1) Although Embodiments 1 and 2 have described the method of controlling bus voltage Vbus to a certain target value, the present invention is applicable to a power controller in any other form. 
       FIG. 19  shows a power controller of a variation. 
     In this power controller, a battery BAT is directly connected to power bus  2 . Although power controller  1  according to Embodiment 1, 2 controls bus voltage Vbus at a certain voltage, power controller  1  shown in  FIG. 19  controls a charging current 
     Ichg of battery BAT at a certain current. 
     A subtractor  131  of an operation processing unit  3 B of the power controller of  FIG. 19  calculates a difference between charging current Ichg, detected by current detector  14 , and current command value Iref and calculates deviation Error. Deviation Error is expressed by Equation (16).
 
Error= Ichg−I ref  (16)
 
     The components of operation processing unit  3 B of the power controller shown in  FIG. 19  except for subtractor  131  can be components similar to those of Embodiment 1 or 2. The power controller of  FIG. 19  can control a charging current of a battery to be constant in accordance with the manner described in the above embodiments. Thus, the power controller can be mounted in, for example, a satellite employing a bus directly connected to a battery and used as a power controller that controls an amount of electric power supplied from a solar battery. 
     (2) Corresponding operations of operation processing unit  3 A and signal generator  4  described in Embodiments 1 and 2 may be configured by hardware or software of a digital circuit. When the functions of operation processing unit  3 A and signal generator  4  are implemented using software, the power controller includes, for example, a processor  1000  and a memory  2000  as shown in  FIG. 20  and can cause processor  1000  to execute a program stored in memory  2000 . 
     (3) Although the term of Vbus is positive and the target value is negative in Equation (1) of a deviation used for describing Embodiment 1, the target value may be positive and the term of Vbus may be negative, and the signs of the subsequent equations may be inverted. 
     (4) Although described here as an example of Equation (2) for the shunt amount which is an amount of operation is PID control using a value obtained by performing proportional, integral, and derivative operations on the difference between battery charging current Ichg and the current command value, the operation amount and the equation may be replaced with any other operation amount and any other equation that can increase an amount of current to be shunted as battery charging current Ichg increases. 
     (5) Although  FIG. 8  shows an example in which a logic circuit is used as a specific implementation means for obtaining a difference ΔP W  of on-pulse width and shunt amount NDUTY_C, a desired result can be obtained by a circuit using an analog time constant or any other method such as an operation using software. 
     (6) Although the values of coefficients C_tm and C_Norm of  FIG. 8  are expressed by Equations (8) to (12), a correction value may be further added in accordance with the actual characteristics of an apparatus. 
     (7) Although carrier wave cw 1  generated by carrier wave generator  401  is a sawtooth wave in the timing chart shown in  FIG. 5  for ease of explanation, an inverse sawtooth wave or a triangular wave of any appropriate duty ratio may be used. 
     (8) Although described here is an example in which signal generator  4  superimposes an offset on common carrier wave cw 1  using carrier wave generator  401  and offset superimposing unit  402 , signal generator  4  may be configured to individually generate carrier waves with different offsets or reduce an offset from an amount of operation. Although signal generator  4  in the present embodiment has been described by taking, as an example, the case in which the number of a switch element serving as a starting point is increased as the switch-on count decreases, similar effects can be achieved even by a manner in which the number of a switch element serving as a starting point is decreased as the switch-on count increases. 
     It is to be understood that the embodiments disclosed herein are presented for the purpose of illustration and non-restrictive in every respect. It is therefore intended that the scope of the present invention is defined by claims, not only by the embodiments described above, and encompasses all modifications and variations equivalent in meaning and scope to the claims. 
     REFERENCE SIGNS LIST 
       1  power controller;  2  power bus;  3 A,  3 B operation processing unit;  4  signal generator;  5  load;  7  sampling unit;  8  timing signal generator;  9  corrector;  10  modulator;  11  offset generator;  12  overvoltage detector;  13  correction value operation unit;  14  current detector;  15 ,  405 ,  551  adder;  16  counter;  17  comparator;  31 ,  131 ,  552 , Sub 1 , Sub 2  subtractor;  32  PID controller;  401  carrier wave generator;  402  offset superimposing unit;  403 ,  557 ,  558 , CmpH, CmpL, Hi 1  to Hin, Lol to Lon comparator;  406  divider;  409  drive signal assignment unit;  410  driving determiner;  411 - 1  to  411 - n  determiner;  553 ,  555 , Mull multiplier;  554 ,  556  integer formatting unit;  1000  processor;  2000  memory; BAT battery; D 1  to Dn diode (backflow prevention element); I 1  to In DC power supply; S 1  to Sn switch element; SL selector; CT 1 , CT 2 ,  559  counter; FF 1 , FF 2 , FF 3  flip-flop; JK, FFd 1  to FFdn JK flip-flop; NOR NOR circuit; ND 1 , ND 2  NAND circuit.