Patent Publication Number: US-2002012146-A1

Title: Synchronizing nodes in an optical communications system utilizing frequency division multiplexing

Description:
BACKGROUND OF THE INVENTION  
       [0001] 1. Field of the Invention  
       [0002] This invention relates generally to optical fiber communications, and more particularly, to the use of independent gain control for different frequency channels in an optical fiber communications systems utilizing frequency division multiplexing.  
       [0003] 2. Description of the Related Art  
       [0004] As the result of continuous advances in technology, particularly in the area of networking, there is an increasing demand for communications bandwidth. For example, the growth of the Internet, home office usage, e-commerce and other broadband services is creating an ever-increasing demand for communications bandwidth. Upcoming widespread deployment of new bandwidth-intensive services, such as xDSL, will only further intensify this demand. Moreover, as data-intensive applications proliferate and data rates for local area networks increase, businesses will also demand higher speed connectivity to the wide area network (WAN) in order to support virtual private networks and high-speed Internet access. Enterprises that currently access the WAN through T1 circuits will require DS-3, OC-3, or equivalent connections in the near future. As a result, the networking infrastructure will be required to accommodate greatly increased traffic.  
       [0005] Optical fiber is a transmission medium that is well suited to meet this increasing demand. Optical fiber has an inherent bandwidth which is much greater than metal-based conductors, such as twisted pair or coaxial cable. There is a significant installed base of optical fibers and protocols such as the SONET protocol have been developed for the transmission of data over optical fibers. The transmitter converts the data to be communicated into an optical form and transmits the resulting optical signal across the optical fiber to the receiver. The receiver recovers the original data from the received optical signal. Recent advances in transmitter and receiver technology have also resulted in improvements, such as increased bandwidth utilization, lower cost systems, and more reliable service.  
       [0006] However, current optical fiber systems also suffer from drawbacks which limit their performance and/or utility. Many of these drawbacks are frequency dependent. For example, optical fibers typically exhibit dispersion, meaning that signals at different frequencies travel at different speeds along the fiber. More importantly, if a signal is made up of components at different frequencies, the components travel at different speeds along the fiber and will arrive at the receiver at different times and/or with different phase shifts. As a result, the components may not recombine correctly at the receiver, thus distorting or degrading the original signal. In fact, at certain frequencies, the dispersive effect may result in destructive interference at the receiver, thus effectively preventing the transmission of signals at these frequencies. Dispersion effects may be compensated by installing special devices along the fiber specifically for this purpose. However, the additional equipment results in additional cost and different compensators will be required for different types and lengths of fiber.  
       [0007] As another example, the electronics in an optical fiber system typically will have a transfer function which is not flat. That is, the electronics will exhibit different gain at different frequencies. In other applications, an electronic equalizer may be used to compensate for these frequency-dependent gain variations in the electronics. However, in an optical fiber system, the electronics produce an electrical signal which eventually is converted to/from an optical form. In order to take advantage of the wide bandwidth of optical fibers, the electrical signal produced by the electronics preferably will have a bandwidth matched to the wide bandwidth of the optical fiber. Hence, any electronic equalizer will also have to operate over a wide bandwidth, which makes equalization difficult and largely impractical.  
       [0008] Furthermore, as optical fiber systems become larger and more complex, there is an increasing need for efficient approaches to manage and control these systems. In a common architecture for optical fiber systems, the system includes a set of interconnected nodes, with data being transmitted from node to node. In these systems, there is commonly also a need for control, administrative or overhead information to be transmitted throughout the system or between nodes. Information describing the overall network configuration, software updates, diagnostic information (including both point to point diagnostics as well as system-wide diagnostics), timing data (such as might be required to implement a global clock if so desired) and performance metrics are just a few examples of these types of information.  
       [0009] Thus, there is a need for optical communications systems which reduce or eliminate the deleterious effects caused by frequency-dependent effects, such as fiber dispersion and the nonflat transfer function of electronics in the system. There is further a need for systems which support the efficient transmission of control and overhead information.  
       SUMMARY OF THE INVENTION  
       [0010] In accordance with the present invention, a method for synchronizing a receiver node with a transmitter node in an optical fiber communications system includes the following steps. At the transmitter node, a reference signal is generated and the transmitter node is synchronized with the reference signal. The reference signal is modulated onto an optical signal, which is transmitted across an optical fiber to the receiver node. At the receiver node, the reference signal is recovered from the optical signal and the receiver node is synchronized with the recovered reference signal. In one embodiment, a harmonic of the reference signal is generated and the harmonic is used to modulate the optical signal. At the receiver node, the harmonic is recovered from the optical signal and then frequency divided to recover the reference signal.  
       [0011] In another aspect of the invention, the reference signal is frequency division multiplexed with a plurality of electrical low-speed channels to form an electrical high-speed channel, which is used to form the optical signal. At the receiver node, the electrical high-speed channel is recovered from the optical signal, and the reference signal is frequency division demultiplexed from the recovered electrical high-speed channel. In one embodiment, the reference signal is located at a frequency lower than that of the electrical low-speed channels.  
       [0012] In yet another aspect of the invention, an optical fiber communications system includes a transmitter node coupled via an optical fiber to a receiver node. The transmitter node includes a local oscillator coupled to an FDM multiplexer. The local oscillator generates a reference signal. The FDM multiplexer combines low-speed channels with the reference signal into an electrical high-speed channel. The receiver node includes an FDM multiplexer, a local oscillator, and electronics coupled to both of the foregoing. The FDM demultiplexer recovers the reference signal from the electrical high-speed channel. The electronics synchronizes the local oscillator with the recovered reference signal. 
     
    
    
     BRIEF DESCRIPTION OF THE DRAWING  
     [0013] The invention has other advantages and features which will be more readily apparent from the following detailed description of the invention and the appended claims, when taken in conjunction with the accompanying drawing, in which:  
     [0014]FIG. 1A is a block diagram of a fiber optic communications system  100  in accordance with the present invention;  
     [0015]FIG. 1B is a block diagram of another fiber optic communications system  101  in accordance with the present invention;  
     [0016]FIG. 2 is a flow diagram illustrating operation of system  100 ;  
     [0017]FIG. 3A- 3 D are frequency diagrams illustrating operation of system  100 ;  
     [0018]FIG. 4A is a block diagram of a preferred embodiment of FDM demultiplexer  225 ;  
     [0019]FIG. 4B is a block diagram of a preferred embodiment of FDM multiplexer  245 ;  
     [0020]FIG. 5A is a block diagram of a preferred embodiment of low-speed output converter  270 ;  
     [0021]FIG. 5B is a block diagram of a preferred embodiment of low-speed input converter  275 ;  
     [0022]FIG. 6A is a block diagram of a preferred embodiment of demodulator  620 ;  
     [0023]FIG. 6B is a block diagram of a preferred embodiment of modulator  640 ;  
     [0024]FIG. 7A is a block diagram of a preferred embodiment of IF down-converter  622 ;  
     [0025]FIG. 7B is a block diagram of a preferred embodiment of IF up-converter  642 ;  
     [0026]FIG. 8A is a block diagram of a preferred embodiment of RF down-converter  624 ;  
     [0027]FIG. 8B is a block diagram of a preferred embodiment of PF up-converter  644 ;  
     [0028]FIG. 8C is a block diagram of another preferred embodiment of RF down-converter  624 ; and  
     [0029]FIG. 8D is a block diagram of another preferred embodiment of RF up-converter  644 ;  
     [0030]FIG. 9A- 9 C are graphs of gain profiles resulting from attenuation due to impairments in a fiber; and  
     [0031]FIG. 9D is a graph illustrating a gain ramp applied to a transmitted signal. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
     [0032]FIG. 1A is a block diagram of a fiber optic communications system  100  in accordance with the present invention. System  100  includes a transmitter  210 B coupled to a receiver  210 A by an optical fiber  104 . Transmitter  210 B and receiver  210 A are both based on frequency division multiplexing (FDM). Transmitter  210 B includes an FDM multiplexer  245  coupled to an E/O converter  240 . The FDM multiplexer  245  combines a plurality of incoming signals  240 B into a single signal using FDM techniques, and E/O converter  240  converts this single signal from electrical to optical form  120 . The E/O converter  240  preferably includes an optical source, such as a laser, and an optical modulator, such as a Mach Zender modulator, which modulates the optical carrier produced by the optical source with an incoming electrical signal. For convenience, the incoming signals  240 B shall be referred to as low-speed channels; the single signal formed by FDM multiplexer  245  as an electrical high-speed channel, and the final optical output  120  as an optical high-speed channel.  
     [0033] Receiver  210 A reverses the function performed by transmitter  210 B, reconstructing the original channels  240 B at the receiver location. More specifically, receiver  120  includes an O/E converter  220  coupled to an FDM demultiplexer  225 . The O/E converter  220 , preferably a detector such as a high-speed PIN diode, converts the incoming optical high-speed channel  120  from optical to electrical form. The frequency division demultiplexer  225  frequency division demultiplexes the electrical high-speed channel into a plurality of low-speed channels  240 A.  
     [0034] The various components in transmitter  210 B and receiver  210 A are controlled by their respective control systems  290 . The control systems  290  preferably also have an external port to allow external control of the transmitter  210 B and receiver  210 A. For example, an external network management system may manage a large fiber network, including a number of transmitters  210 B and receivers  210 A. Alternately, a technician may connect a craft terminal to the external port to allow local control of transmitter  210 B or receiver  210 A, as may be desirable during troubleshooting.  
     [0035] Various aspects of the invention will be illustrated using the example system  100 . However, the invention is not limited to this particular system  100 . For example, FIG. 1B is a block diagram of another fiber optic communications system  101  also in accordance with the present invention. System  101  includes two nodes  110 A and  110 B, each of which includes a transmitter  210 B and receiver  210 A. The two nodes  110  are coupled to each other by two fibers  104 A and  104 B, each of which carries traffic from one node  110  to the other  110 . Fiber  104 A carries traffic from transmitter  210 B(A) to receiver  210 A(B); whereas fiber  104 B carries traffic from transmitter  210 B(B) to receiver  210 A(A). In a preferred embodiment, the fibers  104  also carry control or other overhead signals between the nodes  110 . In an alternate embodiment, the nodes  110  may be connected by a single fiber  104  which carries bidirectional traffic. In other embodiments, the nodes  110  may contain additional functionality, such as add-drop functionality, thus allowing the nodes  110  to from more complex network configurations.  
     [0036]FIG. 2 is a flow diagram illustrating operation of system  100 . At a high level, transmitter  210 B combines low-speed channels  240 B into an optical high-speed channel  120  using FDM techniques (steps  318 B,  316 B and  314 B). As part of this process, the power of each low-speed channel  240 B is adjusted to compensate for estimated gain effects which the low-speed channel  240 B will experience while propagating through system  100  (steps  321  and  323 ). The gain-compensated high-speed channel  120  is then transmitted across fiber  104  (steps  312 ). Receiver  210 A then demultiplexes the received optical high-speed channel  120  into its constituent low-speed channels  240 A (steps  314 A,  316 A and  318 A).  
     [0037] In more detail, low-speed channels  240 B are received  318 B by transmitter  210 B. The FDM multiplexer  245  combines these channels into a high-speed channel using frequency division multiplexing  316 B techniques. Typically, each low-speed channel  240 B is modulated on a carrier frequency distinct from all other carrier frequencies and these modulated carriers are then combined to form a single electrical high-speed channel, typically an RF signal. E/O converter  240  converts  314 B the electrical high-speed channel to optical form, preferably via an optical modulator which modulates an optical carrier with the electrical high-speed channel. The optical high-speed channel  120  is transmitted  312 B across fiber  104  to receiver  210 A.  
     [0038] FIGS.  3 A- 3 C are frequency diagrams illustrating the mapping of low-speed channels  240 B to optical high-speed channel  120  in system  100 . These diagrams are based on an example in which high-speed channel  120  carries 10 billion bits per second (Gbps), which is equivalent in data capacity to an OC-192 data stream. Each low-speed channel  240  is an electrical signal which has a data rate of 155 million bits per second (Mbps) and is similar to an STS-3 signal. This allows 64 low-speed channels  240  to be included in each high-speed channel  120 . The invention, however, is not to be limited by this example.  
     [0039]FIG. 3A depicts the frequency spectrum  310  of one low-speed channel  240 B after pre-processing. As mentioned previously, each low-speed channel  240 B has a data rate of  155  Mbps. In this example, the low-speed channel  240 B has been pre-processed to produce a spectrally efficient waveform (i.e., a narrow spectrum), as will be described below. The resulting spectrum  310  has a width of approximately 72 MHz with low sidelobes. FIG. 3B is the frequency spectrum  320  of the electrical high-speed channel produced by FDM multiplexer  245 . Each of the 64 low-speed channels  240 B is allocated a different frequency band and then frequency-shifted to that band. The signals are combined, resulting in the 64-lobed waveform  320 . FIG. 3C illustrates the spectra  330  of the optical high-speed channel  120 . The RF waveform  320  of FIG. 3B is intensity modulated. The result is a double sideband signal with a central optical carrier  340 . Each sideband  350  has the same width as the RF waveform  320 , resulting in a total bandwidth of approximately 11 GHz.  
     [0040] Receiver  210 A reverses the functionality of transmitter  210 B. The optical high-speed channel  120  is received  312 A by the high-speed receiver  210 A. O/E converter  220  converts  314 A the optical high-speed channel  120 A to an electrical high-speed channel, typically an RF signal. This electrical high-speed channel includes a number of low-speed channels which were combined by frequency division multiplexing. FDM demultiplexer  225  frequency division demultiplexes  316 A the high-speed signal to recover the low-speed channels  240 A, which are then transmitted  318 A to other destinations. The frequency spectrum of signals as they propagate through receiver  210 A generally is the reverse of that shown in FIG. 3.  
     [0041] Note that each low-speed channel  240  has been allocated a different frequency band for transmission from transmitter  210 B to receiver  210 A. For example, referring again to FIG. 3, the low frequency channel  310 A may enter transmitter  210 B at or near baseband. FDM multiplexer  245  upshifts this channel  310 A to a frequency of approximately 900 MHz. E/O converter  240  then intensity modulates this channel, resulting in two sidelobes  350 A which are 900 MHz displaced from the optical carrier  340 . Low-speed channel  310 A propagates across fiber  104  at these particular frequencies and is then downshifted accordingly by receiver  210 A. In contrast, the high frequency channel  310 N is upshifted by FDM multiplexer  245  to a frequency of approximately 5436 MHz and sidelobes  350 N are correspondingly displaced with respect to optical carrier  340 .  
     [0042] In a preferred embodiment, the optical signal carries signals in addition to the sidelobes  350  carrying the low-speed channels  330 . FIG. 3D is the frequency spectrum of an electrical high-speed channel which also includes a pilot tone  328  and a frequency band  326  used for control or other overhead information. For convenience, frequency band  326  shall be referred to as a control channel, although it may carry overhead information other than control signals or be used for purposes other than control.  
     [0043] In general, the control channel  326  provides a communications link between the nodes along the same media (i.e., fiber  104 ) used by the data-carrying sidelobes  350 . The control channel  326  has many uses. For example, the control channel may be used for remote monitoring; performance metrics measured at one node may be communicated to another node or to a central location via the control channel. The control channel may also be used to send commands to each node, for example to set or alter the configuration of a node. When a node first comes onto a network or returns to the network after a fault, the control channel may be used to implement part of the procedure for bringing the node onto the network. For example, the control channel may be established before the data-carrying channels and may then be used to help set up the data-carrying channels. Alternately, the control channel may also be used to establish handshaking between nodes. As a final example, in fault situations, the control channel may be used to gather diagnostic information for fault isolation and also to aid in fault recovery.  
     [0044] The pilot tone  328  is used to synchronize local oscillators used in the transmitter  210 B and receiver  210 A. The transmitter  210 B generates a reference signal at a frequency of 36 MHz and RF electronics at transmitter  210 B are locked to this reference signal. Electronics also generate the pilot tone  328  from the reference signal. In this particular case, the pilot tone  328  is at a frequency of 324 MHz, or the ninth harmonic of the base frequency of 36 MHz. Conventional intensity modulation results in double sideband modulation. The ninth harmonic is used in order to provide adequate separation between the pilot tones  328  and the optical carrier in the final optical signal. At the receiver  210 A, the pilot tone  328  is recovered and frequency divided by nine to recover the original 36 MHz reference signal. Local oscillators at receiver  210 A are locked to the recovered reference signal and local oscillators at transmitter  210 B are locked to the original reference signal. Thus, local oscillators at the receiver  210 A and the transmitter  210 B are locked to each other.  
     [0045] In this embodiment, the control channel  326  has a width of 26 MHz and a center frequency of 816 MHz. The control channel  326  is described in more detail below. In this embodiment, both the control channel  326  and the pilot tone  328  are located at frequencies lower than the data-carrying sidelobes  310 . However, this is not required. Alternate embodiments can locate the control channel(s) and pilot tone(s) at different frequencies, including interspersed among the sidelobes  310  and/or at frequencies higher than the sidelobes  310 .  
     [0046] Since each low-speed channel  240  is allocated a different frequency band, each channel will typically experience a different gain as it propagates through system  100 . For example, fiber losses, such as due to chromatic dispersion or polarization mode dispersion, typically will be different for sidelobes  350 A and  350 N since they are located at different frequencies. Similarly, the gain due to propagation through the various electronic components may also differ since electronics may exhibit different responses at different frequencies. The term “gain” is used here to refer to both losses and amplification.  
     [0047] However, since the frequency band of each low-speed -channel  240  is known, the gain which the low-speed channel  240  will experience as it propagates through system  100  may be estimated  323  and then compensated for  321  by adjusting the power of each low-speed channel. For example, if sidelobe  350 N is expected to experience more loss than sidelobe  350 A due to chromatic dispersion, then sidelobe  350 N may be amplified with respect to sidelobe  350 A in order to compensate for the expected higher loss. The amplification may be applied directly to sidelobe  350 N or at other locations within system  100 , for example to lobe  310 N exiting the FDM multiplexer  245  or to the corresponding low-speed channel  240 B as it enters the system  100 .  
     [0048] The gain may be estimated in any number of ways. For example, with respect to fiber  104 , in one embodiment, standard analytical models are used to estimate the gain due to propagation through fiber  104  at different frequencies due to different physical phenomena. Often, these gain estimates will depend on the length of fiber  104 , which itself may be estimated based on the expected application. Alternately, the length may be measured, for example by using time-domain reflectometry. In a preferred embodiment, a test signal is sent from node  110 A over fiber  104 A to node  110 B. Node  110 B receives the signal and then returns it to node  110 A via fiber  104 B A timer circuit measures the round-trip elapsed time, which is used to estimate the fiber length.  
     [0049] Similarly, the gain estimates for fiber  104  may alternately be determined empirically by measuring the actual gain experienced at different frequencies or by using empirical models. Analogous techniques may be applied to the rest of system  100 . For example, the gain of electronics may be estimated based on models or may be measured by calibrations, for example performed by the manufacturer at the time of production.  
     [0050] FIGS.  9 A- 9 C are graphs illustrating the attenuation resulting from chromatic dispersion. These graphs plot gain, so increased attenuation is shown as low values of gain. Generally speaking, in optical systems using double-sideband optical signals, the attenuation of the detected signal which results from chromatic dispersion is a function of the length of the fiber, denoted by 1, and the frequency of the sidelobe  350  of interest, denoted by f. As shown in FIG. 9A, for a given frequency f, chromatic dispersion results in an increasing attenuation with increasing length  1 , until a null is reached. After a null is reached, the attenuation decreases. Similarly, as shown in FIG. 9B, for a given length of fiber  1 , the attenuation due to chromatic dispersion increases with increasing frequency f, until a null is reached. Then, the attenuation decreases. If the fiber length  1  and frequencies f of the sidelobes  350  are selected so that a null is not reached, then the chromatic dispersion typically results in a gain rolloff with frequency in the detected signal, as shown in FIG. 9C. Polarization mode dispersion generally has a similar behavior.  
     [0051] Thus, if all of the sidelobes  350  were of equal power when they entered a fiber  104  with the gain profile shown in FIG. 9C, the higher frequency sidelobes typically would experience more attenuation in the detected signal as the optical signal propagates through the fiber. This would result in a rolloff in power received at the receiver  210 A at the higher frequencies. Since it is desirable for power for all sidelobes  350  to be roughly equal at the receiver  210 A, it is desirable to compensate for this rolloff effect. Accordingly, at the transmitter  210 B, the power of the higher frequency low-speed channels  240  is boosted  321  with respect to the lower frequency channels  240  so that after propagation through fiber  104 , the sidelobes  350  are of roughly equal power when they reach the receiver  210 A. FIG. 9D is a graph of the gain G applied to compensate for the rolloff. As the inverse of gain g in FIG. 9C (i.e., G=1/ g), the gain G in FIG. 9D increases with increasing frequency and is concave up. This gain profile is also known as a gain ramp. The gain G is shown as a continuous curve. However, in a preferred embodiment, a constant gain is applied across each sidelobe  350 . For example, the gain G at the center frequency of a specific sidelobe  350  may be applied to the entire sidelobe.  
     [0052] When more than one effect is present, the gain G preferably compensates for all significant effects. For example, in some situations, both chromatic dispersion and polarization mode dispersion result in substantial attenuation of the signal. In one embodiment, the compensatory gain function G(f) is determined according to G(f)=G CD (f) G PMD (f), where G CD (f) compensates for attenuation due to chromatic dispersion and G PMD (f) compensates for attenuation due to polarization mode dispersion. In one embodiment, the function G PMD (ƒ) is selected to accommodate for the peak instantaneous differential group delay intended to be tolerated. In a preferred embodiment, the gain G PMD (ƒ) compensates for a peak differential group delay of 46 ps and results in a 3 dB gain applied to low-speed channel number  64 , centered at frequency f=5436 MHz. This 3 dB gain offsets the differential group delay of 46 ps and ensures that data channel  64  arrives with the same power as a data channel propagating without substantial PMD and therefore without a gain ramp. Continuing this example, an instantaneous differential group delay of 70 ps due to polarization mode dispersion results in an optical power penalty of 3 dB.  
     [0053] Other compensatory gain functions G will be apparent. For example, the external optical modulator in E/O converter  240  may result in a rolloff with frequency. The gain G can be used to compensate for this rolloff, for example by using a power amplifier to apply gain to the RF signal entering the modulator.  
     [0054] The gain may also be estimated using closed loop techniques. In other words, the low-speed channel  240  is transmitted across system  100  and a feedback signal is produced responsive to this transmission. The power of the low-speed channel is then adjusted  321  responsive to the feedback signal. As examples, in one embodiment, the feedback signal may depend on the power of the low-speed channel after it has been transmitted across system  100 . In another embodiment, it may depend on the signal to noise ratio or various error rates in the received low-speed channel  240 A.  
     [0055] In a preferred embodiment, the feedback signal is generated by monitor circuitry coupled to the FDM demuliplexer  225  and fed back from receiver  210 A to transmitter  210 B via fiber  104 , as opposed to some other communications channel. In system  101  of FIG. 1B, the control systems  290  may communicate with each other via the bidirectional traffic on these fibers  104 . For example, consider traffic flow from transmitter  210 B(A) across fiber  104 A to receiver  210 A(B). The feedback signal generated at receiver  210 A(B) for this traffic is fed back to transmitter  210 B(A) via the other fiber  104 B. The control system  290  for node  110 A then generates the appropriate control signals to adjust the powers of the low-speed channels. Similarly, the feedback signal for traffic flowing from transmitter  210 B(B) across fiber  104 B to receiver  210 A(A) may be fed back to transmitter  210 B(B) via the other fiber  104 A.  
     [0056] In a preferred embodiment, a frequency band located between the sidebands  350  (see FIG. 3C) and the optical carrier  340  is allocated for control and/or administrative purposes (e.g., for downloading software updates). In a preferred embodiment, this control channel is also used to transmit the feedback signal between the nodes  110  and for time domain reflectometry in order to estimate the length of the fiber. Since it is often desirable to establish initial communications between nodes  110  using the control channel before establishing the actual data links using sidebands  350 , the control channel preferably has a lower data rate and is less susceptible to transmission impairments than the data carrying sidebands  350 . In an alternate embodiment, one of the frequency bands within the electrical high-speed channel  320  is used for the feedback signal.  
     [0057] Referring now to FIG. 3D, in one embodiment, the control channel  326  has a spectral bandwidth of 26 MHz and utilizes alternate mark inversion/frequency-shift keying (AMI/FSK) modulation with a peak frequency deviation of 9 MHz. Data is transmitted at a rate of 2.048 Mbps using the E 1  protocol. Because the control channel  326  transmits at the E 1  data rate, which is lower than the transmission rate of the data-carrying sidebands  310 , control channel  326  is more robust than the data channels  310  and can tolerate lower SNR. Furthermore, because of the lower data rate and because, in the optical signal, the control channel  326  is closer to the optical carrier than the data-carrying channels  350 , the control channel  326  is generally more resistant to fiber impairments than the data channels  350 . Thus, in situations when the data channels  350  are not transmitting properly, the control channel may still be functioning normally. The control channel  326  can then be used by control system  290  to communicate between nodes  110 A and  110 B in order to bring the data channels  350  to normal operation. This situation may occur if there is a fault in the system or upon start up of the system. The control channel  326  can also be used to exchange information during routine operation, as described above.  
     [0058] Any number of techniques may be used to adjust  321  the power of the low-speed channels  240 . For example, if a closed loop technique is used, standard control algorithms such as proportional control may be used. In another approach, a common mode and a differential mode adjustment may be used alternately. In the differential mode adjustment, the total power of all low-speed channels is kept constant while the allocation of power among the various channels is adjusted. Thus, for example, the gain applied to sidelobe  350 A may be increased by a certain amount if the gain applied to sidelobe  350 N is reduced by the same amount, so that the total power in all sidelobes  350  remains constant. In the common mode adjustment, the allocation of power among the various low-speed channels  240  remains constant while the total power is adjusted. Thus, for example, the gain applied to sidelobes  350 A,  350 N and all other sidelobes  350  may be increased by the same amount, thus increasing the total power.  
     [0059] The use of frequency division multiplexing in system  100  allows the transport of a large number of low-speed channels  240  over a single fiber  104  in a spectrally-efficient manner. It also reduces the cost of system  100  since the bulk of the processing performed by system  100  is performed on low-speed electrical signals. In addition, since each low-speed channel is allocated a specific frequency band, the use of frequency division multiplexing allows different gain to be applied to each low-speed channel in an efficient manner, thus compensating for the specific gain to be experienced by the low-speed channel as it propagates through system  100 .  
     [0060] FIGS.  4 - 8  are more detailed block diagrams illustrating various portions of a preferred embodiment of system  100 . Each of these figures includes a part A and a part B, which correspond to the receiver  210 A and transmitter  210 B, respectively. These figures will be explained by working along the transmitter  210 B from the incoming low-speed channels  240 B to the outgoing high-speed channel  120 , first describing the component in the transmitter  120 B (i.e., part B of each figure) and then describing the corresponding components in the  120 A (i.e., part A of each figure). These figures are based on the same example as FIG. 3, namely 64 STS-3 data rate low-speed channels  240  are multiplexed into a single optical high-speed channel  120 . However, the invention is not to be limited by this example or to the specific structures disclosed.  
     [0061]FIG. 4B is a block diagram of a preferred embodiment of transmitter  210 B. In addition to FDM multiplexer  245  and E/O converter  240 , this transmitter  210 B also includes a low-speed input converter  275  coupled to the FDM multiplexer  245 . FDM multiplexer  245  includes a modulator  640 , IF up-converter  642 , and RF up-converter  644  coupled in series. FIGS.  6 B- 8 B show further details of each of these respective components. Similarly, FIG. 4A is a block diagram of a preferred embodiment of receiver  210 A. In addition to O/E converter  220  and FDM demultiplexer  225 , this receiver  210 A also includes a low-speed output converter  270  coupled to the FDM demultiplexer  225 . FDM demultiplexer  225  includes an RF down-converter  624 , IF down-converter  622 , and demodulator  620  coupled in-series, with FIGS.  6 A- 8 A showing the corresponding details.  
     [0062] FIGS.  5 A- 5 B are block diagrams of one type of low-speed converter  270 , 275 . In the transmit direction, low-speed input converter  275  converts tributaries  160 B to low-speed channels  240 B, which have the same data rate as STS-3 signals in this embodiment. The structure of converter  275  depends on the format of the incoming tributary  160 B. For example, if tributary  160 B is an STS-3 signal then no conversion is required. If it is an OC-3 signal, then converter  275  will perform an optical to electrical conversion.  
     [0063]FIG. 5B is a converter  275  for an OC-12 tributary. Converter  275  includes an O/E converter  510 , CDR  512 , TDM demultiplexer  514 , and parallel to serial converter  516  coupled in series. The O/E converter  510  converts the incoming OC-12 tributary  160 B from optical to electrical form, producing the corresponding STS-12 signal. CDR  512  performs clock and data recovery of the STS-12 signal and also determines framing for the signal. CDR  512  also converts the incoming bit stream into a byte stream. The output of CDR  512  is byte-wide, as indicated by the “×8.” Demultiplexer  514  receives the signal from CDR  512  one byte at a time and byte demultiplexes the recovered STS-12 signal using time division demultiplexing (TDM) techniques. The result is four separate byte-wide signals, as indicated by the “4×8,” each of which is equivalent in data rate to an STS-3 signal and with the corresponding framing. Converter  516  also converts each byte-wide signal into a serial signal at eight times the data rate, with the resulting output being four low-speed channels  240 B, each at a data rate of 155 Mbps.  
     [0064] Low-speed input converter  270  of FIG. SA implements the reverse functionality of converter  275 , converting four 155 Mbps low-speed channels  240 A into a single outgoing OC-12 tributary  160 A. In particular, converter  270  includes CDR  528 , FIFO  526 , TDM multiplexer  524 , parallel to serial converter  522 , and E/O converter  520  coupled in series. CDR  528  performs clock and data recovery of each of the four incoming low-speed channels  240 A, determines framing for the channels, and converts the channels from serial to byte-wide parallel. The result is four byte-wide signals entering FIFO  526 . FIFO  526  is a buffer which is used to synchronize the four signals in preparation for combining them into a single STS-12 signal. Multiplexer  524  performs the actual combination using TDM, on a byte level, to produce a single byte-wide signal equivalent in data capacity to an STS-12 signal. Parallel to serial converter  522  adds STS-12 framing to complete the STS-12 signal and converts the signal from byte-wide parallel to serial. E/O converter converts the STS-12 signal to electrical form, producing the outgoing OC-12 tributary  160 A.  
     [0065] Converters  270  and  275  have been described in the context of OC-3 and OC-12 tributaries and low-speed channels with the same date rate as STS-3 signals, but the invention is not limited to these protocols. Alternate embodiments can vary the number, bit rate, format, and protocol of some or all of these tributaries  160 . One advantage of the FDM approach illustrated in system  100  is that the system architecture is generally independent of these parameters. For example, the tributaries  160  can comprise four 2.5 Gbps data streams, 16 622 Mbps data streams, 64 155 Mbps data streams, 192 51.84 Mbps data streams, or any other bit rate or combinations of bit rates, without requiring major changes to the architecture of system  100 .  
     [0066] In one embodiment, the tributaries  160  are at data rates which are not multiples of the STS-3 data rate. In one variant, low-speed input converter  275  demultiplexes the incoming tributary  160 B into some number of parallel data streams and then stuffs null data into each resulting stream such that each stream has an STS-3 data rate. For example, if tributary  160 B has a data rate of 300 Mbps, converter  275  may demultiplex the tributary into four 75 Mbps streams. Each stream is then stuffed with null data to give four 155 Mbps low-speed channels. In another variant, the speed of the rest of system  100  (specifically the modulator  640  and demodulator  620  of FIG. 4) may be adjusted to match that of the tributary  160 . Low-speed output converter  270  typically will reverse the functionality of low-speed input converter  275 .  
     [0067] Referring to FIG. 6B, modulator  640  modulates the 64 incoming low-speed channels  240 B to produced 64 QAM-modulated channels which are input to the IF up-converter  642 . For convenience, the QAM-modulated channels shall be referred to as IF channels because they are inputs to the IF up-converter  642 . In this embodiment, each low-speed channel  240  is modulated separately to produce a single IF channel and FIG. 6B depicts the portion of modulator  640  which modulates one IF channel. Modulator  640  in its entirety would include 64 of the portions shown in FIG. 6B. For convenience, the single channel shown in FIG. 6B shall also be referred to as a modulator  640 . Modulator  640  includes a FIFO  701 , Reed-Solomon encoder  702 , an interleaver  704 , a trellis encoder  706 , a digital filter  708  and a D/A converter  710  coupled in series. Modulator  640  also includes a synchronizer  712  coupled between the incoming low-speed channel  240 B and the filter  708 .  
     [0068] Modulator  640  operates as follows. FIFO  701  buffers the incoming low-speed channel. Reed-Solomon encoder  702  encodes the low-speed channel  240 B according to a Reed-Solomon code. Programmable Reed-Solomon codes are preferred for maintaining very low BER (typ. lower than 10 −12 ) with low overhead (typ. less than 10%). This is particularly relevant for optical fiber systems because they generally require low bit error rates (BER) and any slight increase of the interference or noise level will cause the BER to exceed the acceptable threshold. For example, a Reed-Solomon code of (204,188) can be applied for an error correction capability of 8 error bytes per every 204 encoded bytes.  
     [0069] The interleaver  704  interleaves the digital data string output by the Reed-Solomon encoder  702 . The interleaving results in more robust error recovery due to the nature of trellis encoder  706 . Specifically, forward error correction (FEC) codes are able to correct only a limited number of mistakes in a given block of data, but convolutional encoders such as trellis encoder  706  and the corresponding decoders tend to cause errors to cluster together. Hence, without interleaving, a block of data which contained a large cluster of errors would be difficult to recover. However, with interleaving, the cluster of errors is distributed over several blocks of data, each of which may be recovered by use of the FEC code. Convolution interleaving of depth  0  is preferred in order to minimize latency.  
     [0070] The trellis encoder  706  applies a QAM modulation, preferably  16  state QAM modulation, to the digital data stream output by the interleaver  704 . The result typically is a complex baseband signal, representing the in-phase and quadrature (I and Q) components of a QAM-modulated signal. Trellis encoder  706  implements the QAM modulation digitally and the resulting QAM modulated signal is digitally filtered by filter  708  in order to reduce unwanted sidelobes and then converted to the analog domain by D/A converter  710 . Synchronizer  712  performs clock recovery on the incoming low-speed channel  240 B in order to synchronize the digital filter  708 . The resulting IF channel is a pair of differential signals, representing the I and Q components of the QAM-modulated signal. In alternate embodiments, the QAM modulation may be implemented using analog techniques.  
     [0071] Referring to FIG. 6A, demodulator  620  reverses the functionality of modulator  640 , recovering a low-speed channel  240 A from an incoming IF channel (i.e., analog I and Q components in this embodiment) received from the IF down-converter  622 . Demodulator  620  includes an A/D converter  720 , digital Nyquist filter  722 , equalizer  724 , trellis decoder  726 , deinterleaver  728 , Reed-Solomon decoder  730  and FIFO  732  coupled in series. Demodulator  620  further includes a synchronizer  734  which forms a loop with Nyquist filter  722  and a rate converter phase-locked loop (PLL)  736  which is coupled between synchronizer  734  and FIFO  732 .  
     [0072] Demodulator  620  operates as FIG. 6A would suggest. The A/D converter  720  converts the incoming IF channel to digital form and Nyquist filter  722 , synchronized by synchronizer  734 , digitally filters the result to reduce unwanted artifacts from the conversion. Equalizer  724  applies equalization to the filtered result, for example to compensate for distortions introduced in the IF signal processing. Trellis decoder  726  converts the I and Q complex signals to a digital stream and deinterleaver  728  reverses the interleaving process. Trellis decoder  726  may also determine the error rate in the decoding process, commonly referred to as the channel error rate, which may then be used to estimate the gain of system  100  as described previously. ReedSolomon decoder  730  reverses the Reed-Solomon encoding, correcting any errors which have occurred. If the code rate used results in a data rate which does not match the rate used by the low-speed channels, FIFO  732  and rate converter PLL  736  transform this rate to the proper data rate.  
     [0073] Referring again to transmitter  210 B, IF up-converter  642  receives the 64 IF channels from modulator  640 . Together, IF up-converter  642  and RF up-converter  644  combine these 64 IF channels into a single RF signal using FDM techniques. In essence, each of the IF channels (or equivalently, each of the  64  low-speed channels  240 B) is allocated a different frequency band within the RF signal. The allocation of frequency bands shall be referred to as the frequency mapping, and, in this embodiment, the IF channels may also be referred to as FDM channels since they are the channels which are FDM multiplexed together. The multiplexing is accomplished in two stages. IF up-converter  642  first combines the 64 IF channels into 8 RF channels, so termed because they are inputs to the RF up-converter  644 . In general, the terms “IF” and “RF” are used throughout as labels rather than, for example, indicating some specific frequency range. RF up-converter  644  them combines the 8 RF channels into the single RF signal, also referred to as the electrical high-speed channel.  
     [0074] Referring to FIG. 7B, IF up-converter  642  includes eight stages (identical in this embodiment, but not necessarily so), each of which combines 8 IF channels into a single RF channel. FIG. 7B depicts one of these stages, which for convenience shall be referred to as an IF up-converter  642 . IF up-converter  642  includes eight frequency shifters and a combiner  812 . Each frequency shifter includes a modulator  804 , a variable gain block  806 , a filter  808 , and a power monitor  810  coupled in series to an input of the combiner  812 .  
     [0075] IF up-converter  642  operates as follows. Modulator  804  receives the IF channel and also receives a carrier at a specific IF frequency (e.g., 1404 MHz for the top frequency shifter in FIG. 7B). Modulator  804  modulates the carrier by the IF channel. The modulated carrier is adjusted in amplitude by variable gain block  806 , which is controlled by the corresponding control system  290 , and bandpass filtered by filter  808 . Power monitor  810  monitors the power of the gain-adjusted and filtered signal, and transmits the power measurements to control system  290 .  
     [0076] In a preferred embodiment, each IF channel has a target power level based on the estimated gain due to transmission through system  100 . Control system  290  adjusts the gain applied by variable gain block  806  so that the actual power level, as measured by power monitor  810 , matches the target power level. The target power level may be determined in any number of ways. For example, the actual power level may be required to fall within a certain power range or be required to always stay above a minimum acceptable power. Alternately, it may be selected to maintain a minimum channel error rate or to maintain a channel error rate within a certain range. In this embodiment, variable gain block  806  implements the step of adjusting  321  the power of each low-speed channel  240 .  
     [0077] In alternate embodiments, the power adjustment may be implemented by other elements at other locations or even at more than one location. For example, one gain block may apply a common mode gain to all low-speed channels, and another series of gain blocks at a different location may apply individual gain to each channel (i.e., differential mode gain). However, applying the gain adjustment at the location of variable gain block  806  has some advantages. For example, if the power were adjusted prior to modulator  804 , where each low-speed channel consists of an I and a Q channel, care would need to be taken to ensure that the same gain was applied to both the I and Q channels in order to prevent distortion of the signal. Alternately, if the power were adjusted after combiner  812 , it typically would be more difficult to adjust the power of each individual low-speed channel since combiner  812  produces a composite signal which includes multiple individual channels.  
     [0078] The inputs to combiner  812  are QAM-modulated IF signal at a specific frequency which have been power-adjusted to compensate for estimated gains in the rest of system  100 . However, each frequency shifter uses a different frequency (e.g., ranging in equal increments from 900 MHz to 1404 MHz in this example) so combiner  812  simply combines the 8 incoming QAM-modulated signal to produce a single signal (i.e., the RF channel) containing the information of all 8 incoming IF channels. In this example, the resulting RF channel covers the frequency range of 864-1440 MHz.  
     [0079] Referring to FIG. 8B, RF up-converter  644  is structured similar to IF up-converter  642  and performs a similar function combining the 8 RF channels received from the IF up-converter  642  just as each IF up-converter combines the 8 IF channels received by it. In more detail, RF up-converter  644  includes eight frequency shifters and a combiner  912 . Each frequency shifter includes a mixer  904 , various gain blocks  906 , and various filter  908  coupled in series to an input of the combiner  912 .  
     [0080] RF up-converter  644  operate as follows. Mixer  904  mixes one of the RF channels with a carrier at a specific RF frequency (e.g., 4032 MHz for the top frequency shifter in FIG. 8B), thus frequency upshifting the RF channel to RF frequencies. Gain blocks  906  and filters  908  are used to implement standard amplitude adjustment and frequency filtering. For example, in FIG. 8B, one filter  908  bandpass filters the incoming RF channel and another bandpass filters the produced RF signal, both filters for suppressing artifacts outside the frequency range of interest. Each frequency shifter uses a different frequency (e.g., ranging in equal increments from 0 to 4032 MHz in this example) so combiner  912  simply combines the 8 incoming RF signals to produce the single electrical high-speed channel containing the information of all 8 incoming RF channels or, equivalently, all 64 IF channels received by IF up-converter  642 . In this example, the electrical high-speed channel covers the frequency range of 864-5472 MHz.  
     [0081] RF down-converter  624  and IF down-converter  622  implement the reverse functionalities, splitting the RF signal into its 8 constituent RF channels and then splitting each RF channel into its 8 constituent IF channels, respectively, thus producing 64 IF channels (i.e., FDM channels) to be received by demodulator  620 .  
     [0082] Referring to FIG. 8A, RF down-converter  624  includes a splitter  920  coupled to eight frequency shifters. Each frequency shifter includes a mixer  924 , various gain blocks  926 , and various filters  928  coupled in series. Splitter  920  splits the incoming electrical high-speed channel into eight different RF signals and each frequency shifter recovers a different constituent RF channel from the RF signal it receives. Mixer  924  mixes the received RF signal with a carrier at a specific RF frequency (e.g., 4032 MHz for the top frequency shifter in FIG. 8A), thus frequency downshifting the RF signal to its original IF range (e.g., 864-1440 MHz). Filter  928  then filters out this specific IF frequency range. Each frequency shifter uses a different RF frequency with mixer  924  and thus recovers a different RF channel. The output of RF down-converter  624  is the  8  constituent RF channels.  
     [0083] IF down-converter  622  of FIG. 7A operates similarly. It includes a splitter  820  and  8  frequency shifters, each including a bandpass filter  822 , variable gain block  823 , demodulator  824 , and power monitor  826 . Splitter  820  splits the incoming RF channel into eight signals, from which each frequency shifter will recover a different constituent IF channel. Filter  822  isolates the frequency band within the RF channel which contains the IF channels of interest. Demodulator  824  recovers the IF channel by mixing with the corresponding IF carrier. The resulting 64 IF channels are input to demodulator  620 .  
     [0084] Variable gain block  823  and power monitor  826  control the power level of the resulting IF channel. In a preferred embodiment, each IF channel is output from IF down-converter  622  at a target power in order to enhance performance of the rest of the receiver  210 A. Power monitor  826  measures the actual power of the IF channel, which is used to adjust the gain applied by variable gain block  823  in order to match the actual and target power levels. As described previously, the actual received power level for each low-speed channel may be used to estimate the gain of system  100 . In IF down-converter  622 , the actual receive power level may be determined by dividing the output target power for each IF channel by the gain applied by variable gain block  823  in order to maintain the output target power. In another approach, the actual receive power level may be directly measured, for example by placing a power monitor where variable gain block  823  is located.  
     [0085]FIGS. 8C and 8D are block diagrams of the RF downconverter  624  and RF upconverter  622 , respectively, which explicitly account for the pilot tone  328  and control channel  326 . The RF downconverter  624  in FIG. 8C is the same as that in FIG. 8A except for the following difference. In FIG. 8C, the splitter  920  splits the incoming signal into ten parts, rather than eight, and the RF downconverter  624  includes two additional signal paths coupled to splitter  920  to process the two additional parts. In this example, each of the additional signal paths includes a filter  928  coupled to a variable gain block  926 . The first signal path with filter  928  centered at 816 MHz recovers the control channel  326  and the second with filter  928  centered at 324 MHz recovers the pilot tone  328 .  
     [0086] The RF upconverter  644  in FIG. 8D is changed in a similar manner. Specifically, in addition to the eight signal paths leading to combiner  912  shown in FIG. 8C, the RF upconverter in FIG. 8D includes two additional signal paths. Each signal path includes a variable gain block  908  coupled in series to a filter  908 . One path is for adding the control channel  326  and the other adds the pilot tone  328 .  
     [0087] A preferred embodiment of method  300  will now be described, with reference to the bidirectional system  101  and the further details given in FIGS.  5 - 8 . In the preferred method, the gain applied to each low-speed channel  240  is adjusted in order to optimize the channel error rate measured at the receiver  210 A. Feedback occurs over fibers  104 . More specifically, gain is applied to each of the low-speed channels  240  via variable gain block  806 . This gain is initially selected based on an open-loop estimate. As data is transmitted from transmitter  210 B(A) over fiber  104 A to receiver  210 A(B), trellis decoder  726  determines the channel error rate at the receiver  210 A(B). The channel error rate is fed back to node  110 A via the control channel on fiber  104 B. In this embodiment, the control channel is a frequency modulated, alternate mark inverted, B8ZS-encoded baseband transmitted at 2 Mbps. The gain applied by variable gain block  806  is adjusted to optimize this channel error rate. One optimization approach alternates between differential mode and common mode adjustments. In the differential mode adjustment, the gain is increased for low-speed channels  240  which have unacceptable channel error rates and decreased for low-speed channels  240  with acceptable channel error rates, while keeping the overall power in all low-speed channels constant. In the common mode adjustment, if the median channel error rate is unacceptable, then the gain for all channels  240  is increased by equal increments until the median channel error rate is acceptable. In alternate embodiments, channel performance can be monitored by metrics other than the channel error rate, for example, received power, signal to noise ratio, or bit error rate.  
     [0088] It should be noted that many other implementations which achieve the same functionality as the devices in FIGS.  5 - 8  will be apparent. For example, referring to FIG. 8B, note that the bottom channel occupies the frequency spectrum from 864-1440 MHz and, therefore, no mixer  904  is required. As another example, note that the next to bottom channel is frequency up shifted from the 864-1440 MHz band to the 1440-2016 MHz. In a preferred approach, this is not accomplished in a single step by mixing with a 576 MHz signal. Rather, the incoming 864-1440 MHz signal is frequency up shifted to a much higher frequency range and then frequency down shifted back to the 1440-2016 MHz range. This avoids unwanted interference from the  1440  MHz end of the original 864-1440 MHz signal. For example, referring to FIG. 7B, in a preferred embodiment, the filters  808  are not required due to the good spectral characteristics of the signals at that point. A similar situation may apply to the other filters shown throughout, or the filtering may be achieved by different filters and/or filters placed in different locations. Similarly, amplification may be achieved by devices other than the various gain blocks shown. In a preferred embodiment, both RF down-converter  624  and RF up-converter  644  do not contain variable gain elements. As one final example, in FIGS.  4 - 8 , some functionality is implemented in the digital domain while other functionality is implemented in the analog domain. This apportionment between digital and analog may be different for other implementations. Other variations will be apparent.  
     [0089] The FDM aspect of preferred embodiment  400  has been described in the context of combining 64 low-speed channels  240  into a single optical high-speed channel  120 . The invention is in no way limited by this example. Different total numbers of channels, different data rates for each channel, different aggregate data rate, and formats and protocols other than the STS/OC protocol are all suitable for the current invention. In fact, one advantage of the FDM approach is that it is easier to accommodate low-speed channels which use different data rates and/or different protocols. In other words, some of the channels  240 B may use data rate A and protocol X; while others may use data rate B and protocol Y, while yet others may use data rate C and protocol Z. In the FDM approach, each of these may be allocated to a different carrier frequency and they can be straightforwardly combined so long as the underlying channels are not so wide as to cause the different carriers to overlap. In contrast, in the TDM approach, each channel is allocated certain time slots and, essentially, will have to be converted to a TDM signal before being combined with the other channels.  
     [0090] Another advantage is lower cost. The FDM operations may be accomplished with low-cost components commonly found in RF communication systems. Additional cost savings are realized since the digital electronics such as modulator  640  and demodulator  620  operate at a relatively low data rate compared to the aggregate data rate. The digital electronics need only operate as fast as the data rate of the individual low-speed channels  240 . This is in contrast to TDM systems, which require a digital clock rate that equals the aggregate transmission rate. For OC-192, which is the data rate equivalent to the high-speed channels  120  in system  100 , this usually requires the use of relatively expensive gallium arsenide integrated circuits instead of silicon.  
     [0091] Moving further along transmitter  210 B, E/O converter  240  preferably includes an optical source and an external optical modulator. Examples of optical sources include solid state lasers and semiconductor lasers. Example external optical modulators include Mach Zehnder modulators and electro-absorptive modulators. The optical source produces an optical carrier, which is modulated by the electrical high-speed channel as the carrier passes through the modulator. The electrical high-speed channel may be predistorted in order to increase the linearity of the overall system. Alternatively, E/O converter  240  may be an internally modulated laser. In this case, the electrical high-speed channel drives the laser, the output of which will be a modulated optical beam (i.e., the optical high-speed channel  120 B).  
     [0092] The wavelength of the optical high-speed channel may be controlled using a number of different techniques. For example, a small portion of the optical carrier may be extracted by a fiber optic splitter, which diverts the signal to a wavelength locker. The wavelength locker generates an error signal when the wavelength of the optical carrier deviates from the desired wavelength. The error signal is used as feedback to adjust the optical source (e.g., adjusting the drive current or the temperature of a laser) in order to lock the optical carrier at the desired wavelength. Other approaches will be apparent.  
     [0093] The counterpart on the receiver  210 A is O/E converter- 220 , which typically includes a detector such as an avalanche photo-diode or PIN-diode. In an alternate approach, O/E converter  220  includes a heterodyne detector. For example, the heterodyne detector may include a local oscillator laser operating at or near the wavelength of the incoming optical high-speed channel  120 A. The incoming optical high-speed channel and the output of the local oscillator laser are combined and the resulting signal is detected by a photodetector. The information in the incoming optical high-speed channel can be recovered from the output of the photodetector. One advantage of heterodyne detection is that the thermal noise of the detector can be overcome and shot noise limited performance can be obtained without the use of fiber amplifiers.  
     [0094] The modularity of the FDM approach also makes the overall system more flexible and scaleable. For example, frequency bands may be allocated to compensate for fiber characteristics. For a 70 km fiber, there is typically a null around 7 GHz. With the FDM approach, this null may be avoided simply by not allocating any of the frequency bands around this null to any low-speed channel  240 . As a variant, each of the frequency bands may be amplified or attenuated independently of the others, for example in order to compensate for the transmission characteristics of that particular frequency band.  
     [0095] Various design tradeoffs are inherent in the design of a specific embodiment of an FDM-based system  100  for use in a particular application. For example, the type of Reed Solomon encoding may be varied or other types of forward error correction codes (or none at all) may be used, depending on the system margin requirements. As another example, in one variation of QAM, the signal lattice is evenly spaced in complex signal space but the total number of states in the QAM constellation is a design parameter which may be varied. The optimal choices of number of states and other design parameters for modulator/demodulator  640 / 620  will depend on the particular application. Furthermore, the modulation may differ on some or all of the low speed channels. For example, some of the channels may use PSK modulation, others may use 16-QAM, others may use 4-QAM, while still others may use an arbitrary complex constellation. The choice of a specific FDM implementation also involves a number of design tradeoffs, such as the choices of intermediate frequencies, whether to implement components in the digital or in the analog domain, and whether to use multiple stages to achieve the multiplexing.  
     [0096] As a numerical example, in one embodiment, a (187,204) Reed-Solomon encoding may be used with a rate ¾16-QAM trellis code. The (187,204) Reed-Solomon encoding transforms 187 bytes of data into 204 bytes of encoded data and the rate ¾16-QAM trellis code transforms 3 bits of information into a single 16-QAM symbol. In this example, a single low-speed channel  240 B, which has a base data rate of 155 Mbps would require a symbol rate of 155 Mbps×(204/187)×(⅓) 56.6 Megasymbols per second. Including an adequate guard band, a typical frequency band would be about 72 MHz to support this symbol rate. Suppose, however, that it is desired to decrease the bandwidth of each frequency band. This could be accomplished by changing the encoding and modulation. For example, a (188,205) Reed-Solomon code with a rate ⅚64-QAM trellis code would require a symbol rate of 155 Mbps×(205/188)×(⅕)=33.9 Megasymbols per second or 43 MHz frequency bands, assuming proportional guard bands. Alternately, if 72 MHz frequency bands were retained, then the data rate could be increased.  
     [0097] As another example, an optical modulator  240  with better linearity will reduce unwanted harmonics and interference, thus increasing the transmission range of system  100 . However, optical modulators with better linearity are also more difficult to design and to produce. Hence, the optimal linearity will depend on the particular application. An example of a system-level tradeoff is the allocation of signal power and gain between the various components. Accordingly, many aspects of the invention have been described in the context of the preferred embodiment of FIGS.  3 - 8  but it should be understood that the invention is not to be limited by this specific embodiment.  
     [0098] It should be noted that the embodiments described above are exemplary only and many other alternatives will be apparent. For example, in the embodiments discussed above, the low-speed channels  240  were combined into an electrical high-speed channel using solely frequency division multiplexing. For example, each of the 64 low-speed channels  240 B was effectively placed on a carrier of a different frequency and these 64 carriers were then effectively combined into a single electrical high-speed channel solely on the basis of different carrier frequencies. This is not meant to imply that the invention is limited solely to frequency division multiplexing to the exclusion of all other approaches for combining signals. In fact, in alternate embodiments, other approaches may be used in conjunction with frequency division multiplexing. For example, in one approach, 64 low-speed channels  240 B may be combined into a single high-speed channel  120  in two stages, only the second of which is based on frequency division multiplexing. In particular, 64 low-speed channels  240 B are divided into 16 groups of 4 channels each. Within each group, the 4 channels are combined into a single signal using 16-QAM (quadrature amplitude modulation). The resulting QAM-modulated signals are frequency-division multiplexed to form the electrical high-speed channel.  
     [0099] As another example, it should be clear that the tributaries  160  may themselves be combinations of signals. For example, some or all of the OC-3/OC-12 tributaries  160  may be the result of combining several lower data rate signals, using either frequency division multiplexing or other techniques. In one approach, time division multiplexing may be used to combine several lower data rate signals into a single OC-3 signal, which serves as a tributary  160 .  
     [0100] As a final example, frequency division multiplexing has been used in all of the preceding examples as the method for combining the low-speed channels  240  into a high-speed channel  120  for transmission across optical fiber  104 . Other approaches could also be used. For example, the low-speed channels  240  could be combined using wavelength division multiplexing, in which the combining of channels occurs in the optical domain rather than in the electrical domain. In this approach, the low-speed channels are optical in form, the optical power of each low-speed channel is adjusted, and the power-adjusted optical low-speed channels are combined using wavelength division multiplexing rather than frequency division multiplexing. Many of the principles described above may also be applied to the wavelength division multiplexing approach. Although the invention has been described in considerable detail with reference to certain preferred embodiments thereof, other embodiments are possible. Therefore, the scope of the appended claims should not be limited to the description of the preferred embodiments contained herein.