Patent Publication Number: US-8971060-B2

Title: Method and apparatus for controlling a switching mode power supply during transition of load conditions to minimize instability

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS 
     This application claims priority to Chinese Patent Application No. 200920163199.4 filed Jul. 22, 2009 by inventors Quanqing Wu, et. al., commonly assigned and incorporated in its entirety by reference herein for all purposes. 
     BACKGROUND OF THE INVENTION 
     Embodiments of the present invention are directed to power supply control circuits and power supply systems. More particularly, embodiments of the present invention provide methods and circuits for controlling a switching mode power supply (SMPS). Merely as an example, the methods and circuits have been applied in controlling an SMPS during a transition of load conditions. But embodiments of the invention have a much wider applicability. 
     Regulated power supplies are indispensable in modern electronics. For example, the power supply in a personal computer often needs to receive power input from various outlets. Desktop and laptop computers often have regulated power supplies on the motherboard to supply power to the CPU, memories, and periphery circuitry. Regulated power supplies are also used in a wide variety of applications, such as home appliances, automobiles, and portable chargers for mobile electronic devices, etc. 
     In general, a power supply can be regulated using a linear regulator or a switching mode controller. A linear regulator maintains the desired output voltage by dissipating excess power. In contrast, a switching mode controller rapidly switches a power transistor on and off with a variable duty cycle or variable frequency and provides an average output that is the desired output voltage. 
     Compared with linear regulators, switching mode power supplies have the advantages of smaller size, higher efficiency and larger output power capability. On the other hand, they also have the disadvantages of greater noise, especially Electromagnetic Interference at the power transistor&#39;s switching frequency or its harmonics. 
     Pulse Width Modulation (PWM) and Pulse Frequency Modulation (PFM) are two control architectures of switching mode power supplies. In recent years, green power supplies are emphasized, which require higher conversion efficiency and lower standby power consumption. In a PWM controlled switching mode power supply, the system can be forced to enter into burst mode in standby conditions to reduce power consumption. In a PFM controlled switching mode power supply, the switching frequency can be reduced in light load conditions. PFM-controlled switching mode power supply exhibits simple control topology and small quiescent current. Therefore, it is suitable for low cost small output power applications such as battery chargers and adapters. 
     Even though widely used, conventional SMPS has many limitations. For example, during transition of different output load conditions, especially during relatively large load changes, the SMPS may exhibit unstable output voltages, as described in more detail below. 
     Therefore, there is a need for techniques that can provide more effective control of a switching mode power supply. 
     BRIEF SUMMARY OF THE INVENTION 
     The present invention provides devices and methods for controlling the output voltage of a switching mode power supply. Merely as an example, the methods and circuits have been applied in controlling an SMPS during a transition of load conditions. 
     In an embodiment, a controller according to the invention includes a detection circuit for detecting a change in the output load condition from a heavy load to a light load and a control circuit that disables certain functional blocks of the controller in response to the detected load change in order to reduce the current drain of the controller. It has been observed that by reducing the current drain of the controller during the light load condition, instability at the supply voltage of the controller can be minimized. In another embodiment, the instability can be reduced by supplying more power to the output. 
     In another embodiment of the present invention, a controller device includes a comparator having an input for receiving a feedback signal and configured to compare the feedback signal with a reference voltage. The result of the comparison is then delayed in a delay circuit, and the delayed control signal is then used to turn on and off a high voltage source in order to increase the charging rate of the power supply of the controller device and to compensate for the current drain when the power switch remains inactive. 
     Some embodiments of the present invention provide a controller for a switched mode power supply (SMPS) equipped with a transformer having a primary side winding, a secondary winding, and an auxiliary winding. The controller has a detection circuit for detecting a transition from a first output load condition to a second output load condition of the SMPS, and a control circuit coupled to the detection circuit and configured to output one or more control signals in response to the detected output load transition. The one or more control signals includes a first control signal for turning on a power switch to cause a current flow in a primary winding of the SMPS and/or one or more second control signals for turning off one or more functional circuit blocks in the controller. In a specific embodiment, the control circuit is configured to turn on the power switch and to turn off one or more functional blocks in response to the detected output load transition. 
     Another embodiment of the present invention provides a device for controlling a switched mode power supply (SMPS) equipped with a transformer having a primary side winding, a secondary winding, and an auxiliary winding. The device includes a detection circuit for detecting a transition from a heavy output load condition to a light output load condition of the SMPS and a control circuit for turning on a power switch to cause a current flow in the primary side winding upon detection of the transition. In a specific embodiment, the detection circuit comprises a comparator for comparing a feedback voltage with a reference voltage. In another embodiment, the feedback voltage is greater than the reference voltage during the transition. 
     In some embodiments of the present invention, a switching mode power supply (SMPS) system includes a transformer with a primary winding coupled to a power switch, a secondary winding for providing a regulated output voltage, and a controller. In an embodiment, the controller has a detection circuit having an input for receiving a feedback signal and configured to detect a change in an output load condition, and a control circuit coupled to the detection circuit and configured to output one or more control signals in response to the detected output load transition. The one or more control signals includes a first control signal for turning on a power switch to cause a current flow in a primary winding of the SMPS and/or one or more second control signals for turning off one or more functional circuit blocks in the controller. 
     The devices and methods according to the present invention can be applied both to a conventional pulse width modulator or a pulse frequency modulator. The circuit and method can also be applied to a continuous current mode or a discontinuous mode of operation. Parts and functions of the present invention include a detector circuit and a logic circuit. In an embodiment of the invention, the detector circuit may preferably be a comparison circuit and the logic circuit may preferably be an AND gate. 
     The devices and methods according to the present invention may preferably be applied to switched mode power supply systems having a transformer that includes a primary winding, a secondary winding, and an auxiliary winding. 
     These and other features and advantages of embodiments of the present invention will be more fully understood and appreciated upon consideration of the detailed description of the preferred implementations of the embodiments, in conjunction with the following drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of a conventional AC/DC switching mode power converter system; 
         FIG. 2  is a timing diagram showing certain waveforms of the conventional AC/DC switching mode power converter system of  FIG. 1 ; 
         FIG. 3  is a simplified functional block diagram of a switching mode power supply including selected circuit blocks of a controller in accordance with a first embodiment of the present invention; 
         FIG. 4  is a simplified block diagram illustrating a switching mode power supply including selected functional blocks of a controller in accordance with a second embodiment of the present invention; 
         FIG. 5  is simplified block diagram of a switching mode power supply including selected functional blocks of a controller for in accordance with a third embodiment of the present invention. 
         FIG. 6  is a simplified functional block diagram illustrating selected blocks of a SMPS controller for a switching mode power supply in accordance with a fourth embodiment of the present invention; 
         FIG. 7  is a functional block diagram of a controller for a switching mode power supply in accordance with a fifth embodiment of the present invention; 
         FIG. 8  is a functional block diagram illustrating selected functional blocks of a controller for a switching mode power supply in accordance with a sixth embodiment of the present invention; and 
         FIG. 9  shows voltage waveforms of a switching mode power supply according to embodiments of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  is a functional block diagram of a conventional AC/DC power converter system  100 . As shown, system  100  includes an electromagnetic interference (EMI) filter  102 , a rectifier  104 , a bypass capacitor  106  that converts the voltage from the AC voltage source  101  to an unregulated DC voltage at node  105 . System  100  further includes a transformer  120  having a primary winding  121 , a secondary winding  122 , and an auxiliary winding  123 . Primary winding  121  is coupled to a switch  125  that is switched on and off in response to a drive signal OUT  165 . Switch  125  produces a pulsating current  108  across primary winding  121  that transfers an magnetic energy to secondary winding  122  and auxiliary winging  123 . On the secondary side, a pulsating current  132  flowing through a rectifier  131  is stored in a capacitor  135  that converts pulsating current  132  into a DC voltage, which is further filtered through an inductor  137  and a capacitor  138  to provide a substantially constant output voltage Vo to an output load  139 . 
     In  FIG. 1 , auxiliary winding  123  supplies power to a controller  160  through a rectifier  127  and a smoothing capacitor  128 . Controller  160  includes an input supply terminal Vcc that receives the unregulated power supply from node  105  through a resistor  110  and a capacitor  112  at start up. Controller  160  also includes an under-voltage lockout (UVLO) block, a low voltage drop out (LDO) block, a dc bias block, etc. The UVLO block is configured to detect the level of supply voltage Vcc and the bias block is configured to generate a second voltage V 1  that is used by some other blocks of controller  160 . The UVLO and bias blocks are coupled to the supply voltage Vcc. During the startup of controller  160 , UVLO block monitors the value of Vcc and prevents controller  160  from generating the internal voltage V 1  if the value of Vcc is less than a startup value that is required to initiate operations. 
     Controller  160  also includes a comparator  166  that compares a current sensing signal  167  with a scaled feedback signal  168 . Controller  160  further includes an oscillator (OSC) block that, together with the output of comparator  166 , provides the switching output signal OUT  165  to power switch  125  via a driver block  174 . Current sense signal  167  senses a current  108  flowing across power switch  125 . A leading edge blanking (LEB) block interposed between the current sense signal CS and the input of comparator  166  blanks any current sensing signals that may have high peak magnitudes at the start-up phase for reaching comparator  166 . Comparator  166  compares the voltage  167  generated by current sense resistor  126  of primary winding  121  and the scaled voltage  168  of an optocoupler transistor. The compared output signal contains error information of the regulated voltage Vo and serves to set the primary current  108  flowing across power switch  125 . 
     A feedback circuit  140  is coupled to the output voltage Vo to produce, together with an optocoupler  155 , a feedback signal  170 . Feedback circuit  140  includes resistors R 10  and R 11  that together form a voltage divider to provide an attenuated voltage of Vo to a shunt regulator  152 . Shunt regulator  152  further includes a capacitor C 8  and a resistor R 12  that form a feedback loop compensation circuit. Shunt regulator  152  together with optocoupler  155  form an isolated feedback circuit to control the primary current  108 . A higher current in the optocoupler output transistor results in an decrease in voltage signal FB at the input of controller  160 , and thereby reducing the peak value of the primary current  108  that then effectively lowering the regulated output voltage Vo. 
     Even though power converter system  100  can be used in some applications, it has many limitations. One of the limitations is that it does not handle sudden load change conditions satisfactorily, as described in more detail below. 
     When output load  139  changes from a heavy load to a light load condition, the power consumption in the secondary winding is reduced dramatically and causes a voltage surge at output voltage Vo. This voltage surge is fed back to controller  160  as a feedback signal FB  170  via optocoupler  155 . Comparator  166  then produces an output control signal to driver logic  174  that in turn reduces the primary current  108  by turning off power switch  125 . 
     As controller  160  turns off power switch  125 , auxiliary winding  123  stops supplying pulsating current  129  to charge capacitors  112  and  128 . Capacitors  112  and  128  are used for providing power input to the VCC pin of controller  160  and are also referred to as the VCC capacitor or the Vcc capacitor. Although controller  160  stops switching power switch  125 , it still consumes power because its internal function blocks continue to drain current. This current drainage of internal function blocks causes the voltage supply Vcc from the VCC pin to fall below the cut-off threshold value of UVLO. 
     When output load  139  changes from a heavy load to a light load condition, output voltage Vo overshoots and saturates shunt regulator  152 . In an example, shunt regulator  152  is an adjustable precision shunt regulator AZ431 of BCD Semiconductor Manufacturing Limited. The saturation of shunt regulator  152  causes the voltage at FB to drop to a very low level that, in turn, will disable driver logic  174  of controller  160 . Consequently, regulated output voltage Vo and supply voltage Vcc keep decreasing. When output voltage Vo returns back to its original target voltage level, the feedback voltage at input FB of controller  160  still remains low because of the large value of C 8  of the frequency compensation circuit. Therefore, power switch  125  remains deactivated and supply voltage Vcc continues to drop below UVLO. 
       FIG. 2  is a timing diagram showing waveforms of supply voltage Vcc, output voltage Vo, switching signal OUT  175  and feedback signal FB  170  discussed above. Before time t 1 , controller  160  operates at a heavy load condition and regulates the value of the output voltage Vo to a target value within the desired range of values. Current  129  charges capacitor  128  through rectifier  127 , and Vcc is above a valid operating threshold of UVLO. Driver logic  174  is operational and generates switching pulses OUT to turn on and off power transistor  125  that further maintains the voltage Vcc and output voltage Vo. The LDO block together with the bias generator block generate a second operating voltage v 1  that is used by the FB feedback circuit and other internal blocks of controller  160 . At t 1 , output load  139  changes from the heavy load to a light load condition. This load transition causes a voltage surge at Vo. The voltage surge is fed back to controller  160  via optocoupler  155 . 
     Controller  160  reacts to the surge as a sign that the output voltage needs to regulate down, and thus disables power switch  125  to reduce primary current  108 . As a result, output voltage Vo and Vcc continue to drop in value. As capacitors  112  and  128  have relatively smaller value than capacitors  135  and  138 , voltage Vcc may decrease at a faster rate than that of Vo. At t 2 , when Vcc drops below a first threshold value Vth 1 , an internal electronic circuit (not shown) charges Vcc slowly back to a threshold value Vth 2 . As shown in  FIG. 1 , the Vcc capacitor is charged through resistor  110  from capacitor  106 , thus Vcc will increase. In  FIG. 2 , Vth 1  is a start up threshold voltage of IC and Vth 2  is a shut down threshold voltage. At t 3 , driver logic  174  is operational again and turns on and off power switch  125  that pumps pulsating current  132  via rectifier  131  to capacitor  135  and filter inductor  137  and capacitor  138 . The surge of output voltage Vo is again fed back to controller  160  through optocoupler  155 , and the process repeats. 
     According to embodiments of the present invention, several remedies are provided to alleviate the problems described above. For example, in order to prevent Vcc from dropping below UVLO when the output operation mode changes from heavy load to light load, a large bypass capacitor at Vcc may be utilized. However, a large bypass capacitor will increase the system startup time and cost. Another alternative solution is to reduce the current drain of controller  160  by detecting a load condition change at the output and by switching off certain functional blocks of the controller to reduce power consumption by the controller. Still another alternative include detecting a load condition change at the output and turning on the power switch to provide more power. 
     The following equations provide relationships between the secondary output power and input current for both the continuous current mode (CCM) and discontinuous current mode (DCM): 
                   P   =       V   AV     *     I   AV               (   1   )                   I     P   ⁢     -     ⁢   PEAK       =         I   O         N   t     ⁡     (     1   -     D   ON       )         +         V   IN     ·     D   ON           f   s     ·   L           ,       D   ON     =           N   t     ⁢     V   O             N   t     ⁢     V   O       +     V   IN         ⁢           ⁢     (   CCM   )                 (   2   )                 I     P   ⁢     -     ⁢   PEAK       =       I   RAMP     =           2   ·     P   O           f   s     ·   L         =           V   IN     ·     D   ON           f   s     ·   L       ⁢           ⁢     (   DCM   )                   (   3   )                   I     P   ⁢     -     ⁢   AV       =           I   O     ·     D   ON           N   t     ⁡     (     1   -     D   ON       )         +         V   IN     ·     D   ON   2         2   ·     f   s     ·   L           ,       D   ON     =           N   t     ⁢     V   O             N   t     ⁢     V   O       +     V   IN         ⁢           ⁢     (   CCM   )                 (   4   )                 I     P   ⁢     -     ⁢   AV       =           I     P   ⁢     -     ⁢   PEAK       2     ·     D   ON       =             2   ⁢     P   O           f   s     ·   L         ·     D   ON       =           V   IN     ·     D   ON   2         2   ·     f   s     ·   L       ⁢           ⁢     (   DCM   )                   (   5   )                 D   ON     =             N   t     ⁢     V   O         V   IN       ·         2   ·   L   ·     f   s         R   L           ⁢           ⁢     (   DCM   )               (   7   )               
where V AV =average V AC ; V IN =1.414 V AV ; I AV =average I AC ; T p-AV =average I AC ; L=inductance of the primary winding; N t =N p /N s  (ratio between primary and secondary windings); f s =switching frequency, P O =V O *I O  (secondary output power), and R L =the output load.
 
     According to some embodiments of the present invention, the instability shown in  FIG. 2  can be reduced by increasing the charging rate of Vcc and/or reducing the current drain of the controller. 
       FIG. 3  is a simplified functional block diagram of a switching mode power supply  300  including selected circuit blocks of a controller  360  in accordance with a first embodiment of the present invention. Some of the circuit blocks are similar to those shown in  FIG. 1  and are omitted here to simplify the drawing. As shown, switching mode power supply  300  includes a transformer having a primary winding  321 , a secondary winding  322 , and an auxiliary winding  323 . Switching mode power supply  300  further includes a controller  360 . In one embodiment, controller  360  is implemented using a very high voltage integrated circuit (VHVIC) technology to allow a direct coupling to the high voltage source such as the primary winding  321 . Controller  360  can derive its internal power supplies from the high voltage input HV. Controller may also receive a power supply from a secondary winding through rectifier  327  and a capacitor  328 . At power-on or start up phase, capacitor  328  is charged through the high voltage source to provide a voltage source to controller  360 . Once Vcc reaches a voltage threshold value, a UVLO &amp; DC BIAS block  331  enables a LDO &amp; Protection circuit block that provides protection and internal voltage supplies to controller  360 . Controller  360  turns on and off a power switch  325  coupled to primary winding  321  to deliver energy to secondary winding for maintaining a target voltage output Vo (not shown). Hardware implementation at the secondary winding side can be similar to the circuitry shown in  FIG. 1  including the shunt regulator  152  and opto-coupler  155 . In one embodiment, controller  360  includes a comparator  310  that compares a feedback signal  370  with a reference voltage V 2 . Comparator  310  produces a logic signal that may be used to disable or enable some or all functional blocks of controller  360 . In an embodiment, the output of comparator is coupled with a logic gate  320 . For example, if feedback signal  370  is below reference voltage V 2 , i.e., if the output voltage Vo at the secondary side is higher than a target output voltage range, the output of comparator  310  is logic low and the output of logic gate  320  is asserted low, thus disables UVLO block  331 , LDO block  332 , driver logic block  330 . In another embodiment, the output of logic gate  320  also disables switch  340  that couples the high voltage of the primary winding to the internal Vcc of the controller  360  and switch  350  that couples an internal voltage V 1  to feedback signal  370  via a resistor R 1 . By disabling certain functional blocks within controller  360 , the drain current of controller will be reduced and the voltage Vcc will drop much slower. 
       FIG. 4  is a simplified block diagram illustrating a switching mode power supply  400  including selected functional blocks of a controller  460  in accordance with a second embodiment of the present invention. It is understood that controller  460  may include circuit blocks described above in connections with  FIGS. 1 and 3 . Switching mode power supply  400  includes a transformer having a primary winding  421 , a secondary winding  422 , and an auxiliary winding  423 . Comparator  410  is coupled with a feedback signal ZCD that is derived of a resistive divider R 3 /R 4  coupled to auxiliary winding  423 . Feedback signal ZCD reflects both the change of the regulated output voltage Vo and input voltage (node  105 ). ZCD can further be scaled by multiplying with an average value  453  of the current sensing voltage CS that reflects the current flowing across power switch  425 . The scaled product  454  indicating the output power is then compared with a reference voltage V 2  at comparator  410 , whose output is coupled to logic circuit  420  for providing control signals. Dependent on the result of the comparison at comparator  410 , these control signals are used to disable some functions of controller  460  to reduce the current drain and to obtain a slower Vcc discharge. For example, one such functions is the LDO &amp; Protection circuit block. In another example, the control signals can also be used to control the feedback path of the FB signal. 
       FIG. 5  is simplified block diagram of a switching mode power supply  500  including selected functional blocks of a controller  560  for in accordance with a third embodiment of the present invention. It is understood that controller  560  may include circuit blocks described above in connections with  FIGS. 1 ,  3 , and  4 . As shown in  FIG. 5 , switching mode power supply  500  includes a transformer having a primary winding  521 , a secondary winding  522 , and an auxiliary winding  523 . Comparator  510  is coupled with a feedback signal FB that is then compared with a reference voltage V 2 . If the voltage of feedback signal FB is lower than reference voltage V 2 , comparator will produce a positive signal  512  at its output. Positive signal  512  is coupled to an AND gate  520  whose other input is coupled to a clock signal  540 . Clock frequency  540  is used to enable a driver logic block  530  to turn on and off a power switch  525 . By turning on and off power switch  525 , the voltage supply Vcc of controller  560  will be maintained above a voltage threshold level above UVLO. In other embodiment, this circuit can be easily adapted to other applications, e.g., to an accelerated buildup of Vcc for the controller at startup. 
     Feedback signal FB is further coupled with a first input of a comparator  534  via a scale circuit k. Comparator  534  has a second input coupled with a current sensing resistor  526  via a LEB circuit  533 . Current sensing resistor produces a voltage at the CS input of controller  560 . In the normal operating mode, comparator  534  compares the voltage CS (after a blanking period at startup) at the current sensing resistor  526  and a scaled feedback voltage kFB to produce an error information. The error information is then used to activate driver logic circuit  430  to turn on and off power switch for regulating the output voltage Vo. 
       FIG. 6  is a simplified functional block diagram illustrating selected blocks of a controller  660  for a switching mode power supply in accordance with a fourth embodiment of the present invention. It is understood that controller  660  may include functional blocks described above, depending on the embodiment. In this embodiment, controller  660  includes a comparator  610  that receives a feedback signal FB and compares the FB signal with a reference voltage V 2 . The result of the comparison is delayed in a delay circuit  620  to produce a delayed control signal  625 . Controller  660  also includes an input HV for receiving a high voltage source that may be an unregulated direct current voltage tapped at a primary winding (not shown). Controller  600  further includes a current source  632  that is controlled by an error amplifier  630 . Error amplifier  630  compares a voltage Vcc with a reference voltage V 1  and produces an error control signal  634 . Error signal  634  is then used to control current source  632 . A switch  640  is interposed between voltage Vcc and current source  632  and is turned on and off by the delayed control signal  625 . As a consequence, the decrease of the Vcc when a surge at the output voltage Vo during a light load condition can be slowed down or even compensated by turning of switch  640 . In an embodiment, switch can be an MOS transistor, a transmission gate, or a semiconductor circuitry. 
       FIG. 7  is a simplified functional block diagram of a switching mode power supply  700  including selected circuit blocks of a controller  760  for in accordance with a fifth embodiment of the present invention. It is understood that controller  760  may include functional blocks described above, depending on the embodiment. As shown in  FIG. 7 , controller  700  includes a comparator  734  that compares a scaled feedback signal kFB with a current sensing signal CS. Comparator  734  is configured to regulate an output voltage Vo to a target value within a desired range of values by turning on and off a power switch  725 . Feedback signal FB is further coupled to a comparator  710  that compares feedback signal FB with a reference voltage V 2 . Based on the result of the comparison, a logic gate  720  may disable some functional blocks of controller  700  to reduce the current drain, in order to slow down the rate of the Vcc voltage drop and thereby minimizing instability at voltage Vcc and surges at the regulated output voltage when the output load changes from a heavy load to a light load. In an exemplary embodiment, LDO and protection block  732  is disabled to reduce the power consumption of controller  760 . It is appreciated that other function blocks may also be disabled to save further power consumption of controller  700  and obtain a slower drop rate of Vcc. 
       FIG. 8  is a functional block diagram illustrating selected functional blocks of a controller  860  for a switching mode power supply in accordance with a sixth embodiment of the present invention. It is understood that controller  860  may include functional blocks described above, depending on the embodiment. In this embodiment, controller  860  includes a comparator  810  that compares a feedback signal with a reference voltage V 2 . Based on the result of the comparison, a gate  820  may disable certain functional blocks of controller  860  according to an oscillation frequency  840 . In an exemplary embodiment, the feedback input can be disabled by turning off a switch  850 . In an embodiment, disabling the feedback input prevents a sink current from flowing across resistor R 1  out of controller  860 . 
       FIG. 9  shows voltage waveforms of a switching mode power supply according to embodiments of the present invention. Before time t 1 , the switching mode power supply is operating with a heavy load at the output. At t 1 , the output load condition changes from the heavy load to a light load. This load change causes a voltage surge at Vo that is fed back to the controller via its FB input. The voltage surge is detected by a voltage surge detector, which, as described in connection with  FIGS. 3-8  above, disables certain functional blocks of the controller to reduce the current drain and slow down the drop rate of Vcc. As a result, the waveforms associated with Vcc, Vo, and the feedback signal FB do not show pronounced instability as in the case of a conventional controller (See  FIG. 2 ). And the supply voltage Vcc remains in a valid range that will not activate the under voltage lockout function of the controller. 
     In view of the achieved improvements provided by the illustrative examples disclosed above, it is evident that embodiments of the present invention not only provide devices and methods to minimize instability of the controller voltage supply and the regulated output voltage, but also can provide circuits and methods to decrease power dissipation of the controller when the output condition changes. 
     While the present invention is described with specific embodiments, it is evident that many alternatives and variations will be apparent to those skilled in the art. For example, the disclosed devices and methods of the present invention may also apply to converters with pulse width modulation or pulse frequency modulation, and they may also apply to many other functional blocks such as over current protection block, over temperature block, and many other functional blocks that are not disclosed above.