Patent Publication Number: US-2009238225-A1

Title: 6K pulse repetition rate and above gas discharge laser system solid state pulse power system improvements

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present application is a divisional of U.S. patent application Ser. No. 11/300,979, entitled 6K PULSE REPETITION RATE AND ABOVE GAS DISCHARGE LASER SYSTEM SOLID STATE PULSE POWER SYSTEM IMPROVEMENTS, filed on Dec. 15, 2005, Attorney Docket No. 2005-0091-02, which is a continuation-in-part of U.S. patent application Ser. No. 11/241,850, entitled GAS DISCHARGE LASER SYSTEM ELECTRODES AND POWER SUPPLY FOR DELIVERING ELECTRICAL ENERGY TO SAME, filed on Sep. 29, 2005, Attorney Docket No. 2005-0051-01, and claims priority to U.S. Patent Application No. 60/733,052, filed on Nov. 2, 2005, the disclosures of which are all hereby incorporated by reference. The present application is related to U.S. Pat. No. 6,690,706, entitled HIGH REP-RATE LASER WITH IMPROVED ELECTRODES, issued to Morton et al. on Feb. 10, 2004, and U.S. Pat. No. 6,882,674, entitled 4 KHZ GAS DISCHARGE LASER SYSTEM, issued to Wittak et al on Apr. 19, 2005; and U.S. Pat. No. 6,442,181, entitled EXTREME REPETITION RATE GAS DISCHARGE LASER, issued to Oliver, et al. on Aug. 27, 2002; and U.S. Pat. No. 5,448,580, entitled AIR AND WATER COOLED MODULATOR, issued to Birx, et al. on Sep. 5, 1995, and U.S. Pat. No. 5,315,611, entitled HIGH AVERAGE POWER MAGNETIC MODULATORS FOR METAL VAPOR LASERS, issued to Ball et al. on May 24, 1994, and U.S. patent application Ser. No. 10/607,407, entitled METHOD AND APPARATUS FOR COOLING MAGNETIC CIRCUIT ELEMENTS, filed on Jun. 25, 2003, published on Dec. 30, 2004, Attorney Docket No. 2003-0051-01; the disclosures of each of which are hereby incorporated by reference. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to gas discharge laser systems operating at very high repetition rates of about 6 kHz and above and requiring certain modifications to solid state pulse power systems for supplying power to the electrodes for creating the gas discharges at very high current/voltage and very high pulse repetition rates. 
     BACKGROUND OF THE INVENTION 
     As shown schematically in  FIG. 1  a magnetic switch pulsed power circuit, basically known in the art (with the exception of certain component parameters and operating parameters modified from the known circuitry for operation at 6 kHz and above pulse repetition rates), e.g., for use in supplying high pulse repetition rate (4 kHz and above) electrical pulses between electrodes in a gas discharge laser system, e.g., a KrF or ArF excimer laser or other excimer lasers, e.g., XeCl, or XeF, or other gas discharge lasers, e.g., CO 2  laser systems). 
     Such a pulsed power circuit may include, as illustrated in  FIG. 1 , e.g., a high voltage power supply  20 , a commutator module  50 , a compression head module  60  and a laser chamber module  80 . The high voltage power supply module  20  can comprise, e.g., for a 4 kHz pulse repetition rate laser, a 600 volt rectifier  22  for, e.g., converting the 480 volt three phase normal plant power from an electrical power AC source  10  to about 600 volt DC. An inverter  24 , e.g., converts the output of the rectifier  22  to, e.g., high frequency 600 volt pulses in the range of 10 kHz to 100 kHz. The frequency and the on period of the inverter  24  can be controlled, e.g., by a HV power supply control board (not shown) in order to provide course regulation of the ultimate output pulse energy of the power supply system  20 , e.g., based upon the output of a voltage monitor comprising, e.g., a voltage divider  44 , e.g., in the commutator module  20 . 
     The output of the inverter  24  can be stepped up to about 800 volts in a step-up transformer  26 . The output of transformer  26  can be converted to 800 volts DC by a rectifier  28 , which can include, e.g., a standard bridge rectifier circuit and a filter capacitor  34 . The power supply module  20  can be used to take the DC output of a source  10 , e.g., to charge, e.g., an 5.1 μF charging capacitor C 0  in the commutator module  50  as directed by a control board (not shown), which can, e.g., control the operation of the power supply module  20  to set this voltage. Set points, e.g., within the HV source  10  or power supply control board(s) (not shown) can be provided by a laser system control board (not shown). In the discussed embodiment, e.g., pulse energy control for the laser system can be provided by regulating the voltage supplied by the set of the power source  10  to the power supply module  20  and the power supply module  20  to C 0    42  in the commutator module  50 . 
     The electrical circuits in commutator module  50  and compression head module  60  may, e.g., serve to amplify the voltage and compress the pulses of electrical energy stored on charging capacitor C 0    42  by the power supply  18 , including the source  10  and power supply module  20  module  20 , e.g., to provide 800-1200 volts to charging capacitor C 0 , which during the charging cycle can be isolated from the down stream circuits, e.g., by a solid state switch  46 , which actually may comprise a plurality, e.g., two or three, solid state switches in parallel, e.g., in order to reduce the current flow through each. 
     The commutator module  40 , which can comprise, e.g., the charging capacitor C 0 , which can be, e.g., a bank of capacitors connected in parallel to provide a total capacitance of, e.g., 5.1 μ.F, along with the voltage divider  44 , in order to, e.g., provide a feedback voltage signal to the HV power source  10  or power supply module  20  control board (not shown) which can be used by control board to limit the charging of charging capacitor C 0    42  to a voltage (so-called “control voltage”), which, e.g., when formed into an electrical pulse and compressed and amplified in the commutator  40  and compression head  50 , can, e.g., produce the desired discharge voltage on a peaking capacitor C p    82  and across electrodes  83 , 84  in the lasing cavity chamber  80 . 
     As is known in the art, e.g., for a laser system operating at around 4 kHz, and also for a laser system operating at around 6 kHz or above, such a circuit  50 ,  60 ,  80  may be utilized to provide pulses in the range of 3 or more Joules and greater than 14,000 volts at pulse rates of 4,000 or more pulses per second. In such a circuit, e.g., at 4 kHz and above, about 160 microseconds may be required for DC power source  10  and power supply module  20  to charge the charging capacitor C 0    42  to, e.g., between about 800-1200 volts. At 6 kHz and above the charging time is reduced to about 100 microseconds, and so forth as pulse repetition rate increases. 
     Charging capacitor C 0    42 , therefore, can, e.g., be fully charged and stable at the desired voltage provided the voltage and current applied to the charging capacitor C 0    42  in the amount of time allowed by the pulse repetition rate can be accomplished. For example, when a signal from a commutator control board (not shown) is provided, e.g., to close the solid state switch  46 , which, e.g., initiates a very fast step of converting the 3 Joules of electrical energy stored on charging capacitor C 0    42  into, e.g., a 14,000 volt or more charge on peaking capacitor C p    82  for creating a discharge across the electrodes,  83 ,  84 , provided the charging capacitor C 0  has been adequately charged within the time allotted by the pulse repetition rate of the laser system. 
     The solid state switch  46  may be, e.g., an IGBT switch, or other suitable fast operating high power solid state switch, e.g., an SCR, GTO, MCT, high power MOSFET, etc. A 600 nH charging inductor L 0    48  can be placed in series with the solid state switch  46  and employed, e.g., to temporarily limit the current through the solid state switch  46  while it closes to discharge the charge stored on charging capacitor C 0    42  onto a first stage capacitor C 1    52  in the commutator module,  50  e.g., forming a first stage of pulse compression in the commutator module  50 . 
     For the first stage of pulse generation and compression, the charge on charging capacitor C 0  can be switched onto a capacitor, e.g., a 5.7 μF capacitor C 1 , e.g., in about 4 μs. A saturable inductor L 1    54  can hold off the voltage on capacitor C 1    52  until the saturable reactor L 1    54  saturates, and then present essentially zero impedance to the current flow from capacitor C 1    52 , e.g., allowing the transfer of charge from capacitor C 1    52  through, e.g., a step up transformer  56 , e.g., a 1:25 step up pulse transformer, in order to charge a capacitor C p-1    62  in the compression head module  60 , with, e.g., a transfer time period of about 400 ns, comprising a second stage of compression. 
     The design of pulse transformer  56  is described in a number of prior patents assigned to the common assignee of this application, including, e.g., U.S. Pat. No. 5,936,988. For example, such a transformer  56  is an extremely efficient pulse transformer, transforming, e.g., a 800 volt 5000 ampere, 400 ns pulse to, e.g., a 20,000 volt, 200 ampere 400 ns pulse, which, e.g., is stored very temporarily on compression head module capacitor C p-1    62 , which may also be, e.g., a bank of capacitors. The compression head module  60  may, e.g., further compress the pulse. A saturable reactor inductor L p-1    64 , which may be, e.g., about a 125 nH saturated inductance, can, e.g., hold off the voltage on capacitor C p-1    62  for approximately 400 ns, in order to, e.g., allow the charge on C p-1    62  to flow, e.g., in about 100 ns, onto a peaking capacitor C p    82 , which may be, e.g., a 10.0 nF capacitor located, e.g., on the top of a laser chamber and which peaking capacitor C p    82  is electrically connected in parallel with the laser system electrodes  83 ,  84 . 
     This transformation of a, e.g., 400 ns long pulse into a, e.g., 100 ns long pulse to charge peaking capacitor C p    82  can make up, e.g., the second and last stage of compression. About 100 ns after the charge begins flowing onto peaking capacitor C p    82  (which may be a bank of capacitors in parallel) mounted on top of and as a part of the laser chamber in the laser chamber module, the voltage on peaking capacitor C p    82  will have reached, e.g., about 20,000 volts and a discharge between the electrodes begins. The discharge may last, e.g., about 50 ns, during which time, e.g., lasing occurs within the resonance chamber of the, e.g., excimer laser. 
     According to aspects of an embodiment of the present invention may comprise operation of laser systems requiring, e.g., precisely controlled electrical potentials in the range of about 12,000 V to 20,000 V be applied between the electrodes at around 6,000 Hz and above (i.e., at intervals of about 166 micro seconds). As indicated above in known magnetic switch pulse power systems the charging capacitor bank C 0    42  can be is charged to a precisely predetermined control voltage and the discharge can be produced by closing the solid state switch  46  which can then allow the energy stored on the charging capacitor C 0    42  to ring through the magnetic compression-amplification circuitry  50 ,  60  and  80  to produce the desired potential across the electrodes  83 ,  84 . The time between the closing of the switch  46  to the completion of the discharge is only a few microseconds, (i.e., about 5 microseconds) but the charging of C 0    42  can require a time interval much longer than 166 microseconds. It is known, however, to reduce the charging time, e.g., by using a larger power supply. Alternatively, using power supplies in parallel can reduce the charging time. For example, it has bee shown that one is able to operate at around 4,000 Hz, e.g., by using three prior art power supplies such as those shown illustratively as element  18  in  FIG. 1 , arranged in parallel. 
     In such an embodiment, one may also utilize the same basic design as in the prior art shown in  FIG. 1  for the portion of the pulse power system downstream of the solid state switch  46 . One may also implement a known different technique for charging C 0    42 , e.g., as illustrated schematically and in block diagram form in  FIGS. 2 and 3 . Applicants&#39; assignee Cymer, Inc. refers to such techniques as resonant charging, of which at least two alternative apparatus and methods may be employed as illustrated by way of example in  FIGS. 2 and 3 , respectively, which are taken from the above referenced U.S. Pat. No. 6,442,181, resulting in, e.g., very fast charging of C 0    42 . 
     A standard dc power supply  200  having a 208 VAC/90 amp input and an 800 VDC 50 amp output may be used. The power supply  200  may be a dc power supply adjustable from approximately 600 volts to 800 volts. The power supply  200  may be attached directly to a storage capacitor C- 1   202 , in a resonant charger  220 , which may be, e.g., a 1033 μF capacitor. When the power supply  200  is enabled it turns on and regulates a constant voltage on the C-1 capacitor  202 . The performance of the system is somewhat independent of the voltage regulation on C- 1   202 . The power supply  200  may be, e.g., a constant current, fixed output voltage power supply such as is available from Elgar, Universal Voltronics, Kaiser and EMI. 
     The power supply  200  may continuously charges the 1033 μ.F capacitor  202  to the voltage level commanded by the control board  204 , in the embodiment of  FIG. 2 . The control board  204  may also command IGBT switch  206  (which may be a plurality of switches in parallel) closed and open to transfer energy from capacitor  202  to capacitor C 0    42 . A charging inductor  208  in the resonant charger  220  may sets up the transfer time constant in conjunction with capacitor  202  and  42  and limits the peak charging current. Control board  204  receives a voltage feedback  212  (e.g., as shown in  FIGS. 2 and 3 ) that is proportional to the voltage on capacitor  42  and a current feedback  214  (e.g., as shown in  FIG. 3  that is proportional to the current flowing through inductor  208 . From these two feedback signals control board  204  can calculate in real time, e.g., a final voltage on capacitor  42  should IGBT switch  206  open at that instant of time. Therefore with a command voltage  210  fed into control board  204  a precise calculation can be made of the stored energy within capacitor  42  and inductor  208  to compare to the required charge voltage commanded  210 . From this calculation, the control board  204  can, e.g., also determine the exact time in the charge cycle to open IGBT switch  206 . 
     After IGBT switch  206  opens the energy stored in the magnetic field of inductor  208  can transfer to capacitor  42  through a free-wheeling diode ( 215  in  FIG. 2  or  217  in  FIG. 3 ). The accuracy of the real time energy calculation can determine, e.g., the amount of fluctuation dither that will exist on the final voltage on capacitor  42 . Due to the extreme charge rate of this system, too much dither may exist to meet a desired systems regulation need of ±0.05%. Therefore the circuit may also include, for example, a de-qing circuit or a bleed-down circuit as discussed below. 
     A second resonant charger system is shown illustratively and in block diagram form by way of example in  FIG. 3 . This circuit is similar to the one shown in  FIG. 2 . Principal circuit elements may include the three-phase power supply  200  with a constant DC current output, the source capacitor C- 1   202  that is an order of magnitude or more larger than the existing C 0  capacitor  42  (e.g., a 1033 μF capacitor. Switches Q 1 ,  206 , Q 2   218 , and Q 3   216  can be used to control current flow for charging and maintaining a regulated voltage on charging capacitor C 0    42 . Diodes D 1   215 , D 2   217 , and D 3   219 , which may be a bank of diodes in parallel, can provide for the direction of current flow. Resisters R 1   230 , and R 2   232  provide a voltage divider circuit for voltage feedback  212  to the control circuitry on the control board  204 . Resistor R 3   240 , shown in  FIG. 2  to be a 0.001 ohm resistor and in  FIG. 3  to be a 500 ohm resistor can be used to allows for rapid discharge of the voltage on the charging capacitor C 0    42  to bleed down the charge on the capacitor  42  in the event of an over charge on the capacitor  42 , as detected, e.g., by the voltage divider circuit of resisters  230 ,  232 . A resonant inductor L 1 ,  242  between capacitors C- 1   202  and C 0    42  may serve to limit current flow and setup charge transfer timing. The control board  204  may provide commands to the switches Q 1   206 , Q 2   218 , and Q 3   216  to, e.g., open and close the switches based upon, e.g., circuit feedback parameters. The difference in the circuit of  FIG. 2  from that of  FIG. 3  is, e.g., the addition of switch Q 2   218  and diode D 3   219 , which can provide a de-Qing function. This switch  218  can be used to improve the regulation of the circuit by allowing the control unit  204  to short out the inductor L 1   208  during the resonant charging process. This “de-Qing” can be used, e.g., to prevent additional energy stored in the charging inductor, L 1   208 , from being transferred to capacitor C 0 . 
     Prior to the need for a laser pulse the voltage on capacitor C- 1   202  can be charged to, e.g., 600-800 volts and switches Q 1   206 , Q 2   218  and Q 3   216  may be open. Upon a command from the laser system controller (not shown), the control board  204  can provide a command  206 ′ to switch Q 1   206  to close the switch Q 1   206 . At this time current would flow from capacitor C- 1   202  to charging capacitor Co  42  through the charging inductor L 1   208 , since switch Q 2  can be open at this time. A calculator  205  on the control board  204  could be used to evaluate the voltage on C 0    42  and the current flowing in inductor L 1   208 , from feedback signals  212 ,  214 , relative to a command voltage set point from the laser. Switch Q 1   206  can then be opened by a command  206 ′ from the control board  204  when the voltage on charging capacitor C 0    42  plus the equivalent energy stored in inductor L 1   206  equals the desired command voltage. The calculation is: 
         V   f   =[V   C0s   2 +(( L   1   *I   L1s   2 )/ C   0 )] 0.5    
     where: V f =a final voltage on C 0  after switch Q 1   206  opens and the current in inductor L 1   208  goes to zero; V C0s  is the starting voltage on C 0  when switch Q 1   206  opens; I L1s  is the current flowing through L 1  when switch Q 1   206  opens. After switch Q 1   206  opens the energy stored in inductor L 1   208  continues transferring to C 0  through diode D 2   217  until the voltage on C 0  approximately equals the command voltage. At this time switch Q 2   218  can be closed and current stops flowing to charging capacitor C 0    42  and is directed through diode D 3   219 . In addition to the “deque” circuit,  218 ,  219 , switch Q 3   216  and resistor R 3   240  form a bleed-down circuit to allow additional fine regulation of the voltage on C 0  to a target charging voltage. 
     Switch Q 3  of bleed down circuit  216 ,  240  can be commanded to close, e.g., by the control board  204 , e.g., when current flowing through inductor L 1   208  stops and the voltage on charging capacitor C 0  can be bled down to the desired charging voltage. Then switch Q 3   216  can be opened. The time constant of capacitor C 0    42  and resistor R 3   240  can be selected to be sufficiently fast to bleed down capacitor C 0    42  to the commanded charging voltage without being an appreciable amount of the total charge cycle. 
     As a result, the resonant charger  220  can be configured with three levels of regulation control. Somewhat crude regulation may be provided by the energy calculator and the timing of the opening of switch Q 1   206  during the charging cycle. As the voltage on C 0  nears the target charging voltage value, the deque switch Q 2   218  may be closed, stopping the resonant charging when the voltage on C 0  is at or slightly above the target value. Finally, as a third control over the voltage regulation the bleed-down circuit of switch Q 3   216  and R 3   240  can be used to discharge C 0  down to the precise target value. 
     According to aspects of an embodiment of the present invention these known magnetic switch pulsed power supply systems may carry out parallel non-resonant charging, e.g., for operation of laser systems at pulse rates of 4,000 Hz to 6,000 Hz can be accomplished with the prior art charging system technology shown as element  20  in  FIG. 1 . However, to provide the needed charging speed, much greater charging capacity is required. For example, applicants&#39; assignee&#39;s laser systems have successfully been operated at laser output pulse (and thus also gas discharge electrical pulse) repetition rates of 4,900 Hz using, e.g., three of the  FIG. 1  power supplies in parallel. For operation at 6,000 Hz five (preferably six) of these power supplies could be needed. 
     According to aspects of an embodiment of the present invention the resonant chargers of  FIGS. 2 and 3  may be employed but applicants have found that certain other modifications and improvements may be necessary for operation at 6 kHz and above. The present application addresses such issues. It will also be understood that commands mentioned above, e.g., to certain switches may be, e.g., in the form of an applied voltage or current to open or shut the respective switch. The solid state switch  46  may comprise an P/N CM 800 HA-34H IGBT switch provided by Powerex, Inc. with offices in Youngwood, Pa. In a preferred embodiment, as noted, at least two such switches may be used in parallel. Inductors, e.g.,  54  and  64  may be saturable inductors similar to those used in prior systems as described in the above referenced U.S. Pat. Nos. 5,448,580 and 5,315,611. It is also discussed in Ness, et al., “A Decade of Solid State Pulsed Power Development at Cymer Inc.” Proceedings of the 26th IEEE International Power Modulator Symposium and High Voltage Workshop, San Francisco, (2004), pp 228-233. 
     A technique for water cooling a step-up transformer is disclosed in U.S. Pat. No. 5,448,580, entitled AIR AND WATER COOLED MODULATOR, issued to Birx, et al on Sep. 5, 1995 disclosing:
         With reference again to  FIG. 5 , the system used in the present invention to cool transformer  22  is also shown. A cold plate  106  is attached to the primary winding assemblies  20  to carry heat therefrom. Cold plate  106  may be cooled, for example, by flowing cooling water through channels  108  in cold plate  106 . In the present embodiment, cooling water is supplied to cold plate  106  using flexible tubing, not shown. (Col. 9, lines 19-26)       

     The referenced  FIG. 5  simply shows a single channel passing through a single piece cooling plate. 
     A jitter control circuit is discussed in Huang, et al., “Low Jitter And Drift High Voltage IGBT Gate Driver, Proceedings of the 14th IEEE Pulsed Power Conference, Dallas (2003), pp 127-130, Abstract No. 100055. 
     SUMMARY OF THE INVENTION 
     A method and apparatus for operating a very high repetition gas discharge laser system magnetic switch pulsed power system is disclosed, which may comprise a solid state switch, a charging power supply electrically connected to one side of the solid state switch; a charging inductor electrically connected to the other side of the solid state switch; a deque circuit electrically in parallel with the solid state switch comprising a deque switch; a peaking capacitor electrically connected to the charging inductor, a peaking capacitor charging control system operative to charge the peaking capacitor by opening the deque switch and leaving the solid state switch open and then shutting the solid state switch. The solid state switch may comprise a plurality of solid state switches electrically in parallel. The peaking capacitor charging control system may be operative to charge the peaking capacitor by leaving the deque switch open until substantially all of the electrical energy stored in the charging inductor has been removed before shutting the solid state switch. The very high repetition gas discharge laser system magnetic switch pulsed power system may comprise a solid state switch; a charging power supply electrically connected to one side of the solid state switch; a charging inductor electrically connected to the other side of the solid state switch; a peaking capacitor electrically connected to the charging inductor, a delay circuit operative to charge the peaking capacitor with electrical energy stored in the charging inductor prior to shutting the solid state switch. The very high repetition gas discharge laser system magnetic switch pulsed power system may comprise a step-up transformer comprising a plurality of winding pucks each comprising a turn primary winding around a secondary winding; each of the plurality of pucks contained in at least two separate sections of primary winding pucks laid out on a step-up transfer mounting board at angles to each other generally forming an L or a U or an O shaped compilation having a first and a second end; a cooling plate having a plurality of sections each respectively in thermal contact with a respective one of the at least two separate sections of the primary winding pucks; the cooling plate may comprise a plurality of cooling channels arranged in at least one grouping of a pair of channels extending in a flow direction from the first end to the second end and returning to the first end, from a cooling fluid inlet at the first end to a cooling fluid outlet at the first end. The cooling channels may comprise a channel internal to the cooling plate. The cooling channel may be formed in at least a first half of the cooling plate and the first half of the cooling plate is joined to a second half of the cooling plate. The cooling channel may comprise a cooling fluid duct contained in a cooling fluid duct passage groove formed in a surface of the cooling plate. The cooling fluid duct may comprise thermally conductive tubing. The very high repetition gas discharge laser system magnetic switch pulsed power system may comprise a step-up transformer comprising a plurality of winding pucks each comprising a turn primary winding around a secondary winding; a void space between an internal surface of each respective primary winding puck and an insulation sleeve on the secondary winding; and insulation fluid in the void space. The insulation fluid may comprise a dielectric gas, e.g., a noble gas, e.g., N 2 , or a dielectric liquid, e.g., a dielectric oil. The very high repetition gas discharge laser system magnetic switch pulsed power system may comprise a solid state switch anti-jitter and anti-drift circuit which may comprise an optoisolator circuit spanning the boundary between the high voltage side of the circuit and the low voltage side of the circuit, which may comprise an opto-transmitter on the low voltage side of the circuit and an opto-receiver on the high voltage side of the circuit. The circuit may comprise a comparator in series with the opto-receiver and the solid state switch between the opto-receiver and the solid state switch. The opto-transmitter may be connected to a trigger input signal and the comparator may be connected to an MOSFET driver circuit. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows schematically and partly in block diagram form a magnetic switch pulsed power supply system useful according to aspects of an embodiment of the present invention; 
         FIG. 2  shows schematically and partly in block diagram form a resonant charging circuit useful according to aspects of an embodiment of the present invention; 
         FIG. 3  shows schematically and partly in block diagram form a resonant charging circuit useful according to aspects of an embodiment of the present invention; 
         FIG. 4  shows illustratively by way of example in schematic and partly in block diagram form a delay circuit according to aspects of an embodiment of the present invention; 
         FIG. 5A  shows schematically and partly in block diagram form a known solid state pulse power supply system solid state switch anti-jitter and drift control circuit; 
         FIG. 5B  shows schematically and partly in block diagram format a solid state pulse power supply system solid state switch anti-jitter and drift control circuit according to aspects of an embodiment of the present invention; 
         FIG. 6  shows a plan view of a portion of a cooling plate for a solid state pulse power supply system step-up transformer according to aspects of an embodiment of the present invention; 
         FIG. 7  shows a perspective view of a cooling plate for a solid state pulse power supply system step-up transformer according to aspects of another embodiment of the present invention; 
         FIG. 8  shows a side view of the embodiment of  FIG. 7 ; 
         FIG. 9  shows a plan view of a solid state pulse power supply system step-up transformer according to aspects of an embodiment of the present invention; 
         FIG. 10  shows a cross sectional side view of a section of a solid state pulse power supply system step-up transformer according to aspects of another embodiment of the present invention, along lines  10 - 10  of  FIG. 9 ; 
         FIG. 11  shows an enlarged view of a portion of the embodiment of  FIG. 10 ; 
         FIG. 12  Shows an orthogonal perspective view of a portion of the embodiment of  FIG. 9 ; 
         FIG. 13  shown an orthogonal perspective view of a primary winding puck of a solid state pulse power supply system step-up transformer according to aspects of an embodiment of the present invention; 
         FIG. 14  shows a perspective view of an end flange for a section of a solid state pulse power supply system step-up transformer according to aspects of an embodiment of the present invention; 
         FIG. 15  shows an orthogonal partially cut away view of the end flange of  FIG. 14 ; 
         FIG. 16  shows a perspective orthogonal view of a puck isolator according to aspects of an embodiment of the present invention; 
         FIG. 17  shows schematically in more detail portions of the prior art pulse power circuit of  FIG. 1 ; and, 
         FIG. 18  shows schematically modifications to the circuit of  FIG. 17  according to aspects of an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     According to aspects of an embodiment of the present invention an issue to address is that the peak current in the charging inductor of a resonant charger (“RC”) module, e.g.,  220  shown illustratively in  FIGS. 2 and 3  can increase with the 2nd (and on) pulses in a burst of laser system pulsed output light beam pulses since current (stored or recovered energy) may already exist in the charging inductor, e.g., inductor L 1   208  shown, e.g., in  FIGS. 2 and 3 , remaining from the energy recovery cycle of the previous pulse. When the charging switch, e.g., switch Q 1   206  shown illustratively in  FIGS. 2 and 3 , is closed, current also flows from the C-1 capacitor  202 , also illustratively shown in  FIGS. 2 and 3 . This current adds to the already existing energy recovery current and can cause the peak current in the switch Q 1   206  to go higher than it would be for the first pulse. This increases the requirements (for current) of the charging switch Q 1   206  (and additional components e.g., D 1   215  and L 1   208 ) and can also lead to higher losses in the charging switch Q 1   206  (and other components e.g., D 1   215  and L 1   208 ). 
     In order to deal with this, applicants have proposed the implementation a circuit, shown schematically and partly in block diagram form in  FIG. 4 , which can lessen the current flow through the switch Q 1   206  and the other components, e.g., by delaying the closure of the charging switch Q 1   206  until the energy recovery current already stored in the charging inductor L 1   208  has had a chance to decay and thereby charge the charging capacitor C 0    42  in the commutator of the pulsed power supply system. The circuit  300  shown illustratively by way of example in  FIG. 4  may, e.g., employ a charging control calculator  205  on the charging control board  204 , e.g., to simply delay the charging switch  206  closure until the current passing through the charging inductor  208  drops below a preset value. 
     In this manner, the operation of the known resonant charger circuit, e.g., as shown in  FIG. 3  is modified, e.g., such that switch  218  in the de-quing circuit  218 ,  219  is opened once the resonant charger circuit is commanded to begin charging, allowing current flowing through the charging inductor L 1   208  to flow into the charging capacitor C 0    42 , until substantially all of the current is used to charge the charging capacitor C 0    42 , i.e., current flow in the charging inductor L 1   208  is zero or substantially zero. Once the current from the prior pulse has been dissipated, the charging switch  206  is commanded closed by the charging control calculator  205  to allow additional charging of the C0 capacitor  42  from the C-1 capacitor  202 . The normal charging sequence is then followed where the charging switch  206  is opened by the energy calculator circuit and then the deque switch  218  and bleed switch  216  are closed to achieve precise regulation of the charging voltage. In this way, the total current seen by the charging switch and other components may be limited to less that what it would be with the dissipation of the current in the charging inductor L 1   208  and the charge on capacitor C- 1   202  through the switch  206  and other components. 
     Turning now to  FIG. 5A  there is shown by way of illustration a of a prior art solid state pulse power system solid state switch drift and jitter control circuit  246  that is currently on the market in laser systems sold by applicant&#39;s assignee, such as XLA  100  multichamber laser systems. The jitter control circuit  246  as discussed in the above noted reference, may comprise, e.g., an IGBT solid state switch, e.g.,  46 , e.g., one as noted above, (which may be one of a plurality of such IGBTs in parallel) having an IGBT gate  248  and an IGBT emitter  249  and a pair of fast switching MOSFETs. An N-channel MOSFET  250  and a P-channel MOSFET  252 . The circuit may also include a MOSFET driver  254 , which may be connected to the gate of N-Channel MOSFET by a resistor  254  and a diode  256  and to the gate of the P-channel MOSFET  252 . 
     The circuit  246  may also comprise an optocoupler  258  connected across the low voltage to high voltage transition of the circuit  246 , the high voltage side being connected to a DC/DC converter  270 , a model THI-2421, made by TRACO ELECTRONIC AG, Switzerland, which can, e.g., convert a DC voltage supplied by a DC power supply  272  to the DC voltage connected, e.g., to the collector of the IGBT  46 , providing a positive rail  274  to negative rail  276  voltage on the emitter  249  of the IGBT  46  when the IGBT switch  46  is shut. 
     The circuit  246  may also comprise a resistor  282 , which may be a 1670 ohm resistor, connected between the positive rail  274  and common ground  249  and two zener diodes  284  in parallel, e.g., a model 1N4734A, made by ON Semiconductor, U.S.A. connected between the negative rail  276  and common ground  249 . The circuit  246  may also comprise a capacitor, e.g., a 100 μF  290  connected to the positive rail  274  and the IGBT emitter  249  and a capacitor, e.g., a 100 μF capacitor. 
     Such a circuit  246 , e.g., with a high speed optocoupler  258 , e.g., a model HCPL-2611#020 which can be obtained from AGILENT TECHNOLOGIES, U.S. A., an ultrafast MOSFET driver  254 , e.g., a model IXDD404PI, which can be obtained from IXYS CORPORATION, U.S.A. and the fast switching MOSFETs, e.g., model IRFU5305, and IRFU4105, which can be obtained from INTERNATIONAL RECTIFIER, U.S.A., can be utilized to insure, e.g., minimum jitter, turn on delay, turn on time, turn off time, turn-off delay, turn on/off drift and power loss from the receipt of a trigger input signal  259  to the shutting of the IGBT  46  and the application of the voltage on charging capacitor C 0    42  onto capacitor C 1    52  through inductor L 0    48 , as illustrated in the circuit of  FIG. 1 . 
     In operation, e.g., the circuit  246  provides a fully isolated gate driver operable up to relatively high C pk  discharge pulse rates, e.g., around 4000 pulses per second, using the high isolation voltage optocoupler  258 , and optoisolator, and the DC/DC converter  270  to isolate the trigger signal  259  from the high voltage side of the circuit  246 . The resistor  282  and zener diode  284  can provide voltage regulation to generate the positive rail  274  and reference ground  259 . The outputs of the N-channel MOSFET  250  and P-channel MOSFET  252  may be connected common drain for rail to rail output to the IGBT gate  248  and emitter  249 , e.g., to ensure reliable operation. The IGBT  46  gate driver  254  may be mounted, close to, e.g., directly on top of the IGBT  46  to minimize inductances. The resistor  254 , which may be, e.g., a 100 ohm resistor, in parallel with a diode, e.g., a Schottky, e.g., a model IN5818, made by ON SEMICONDUCTOR, U.S.A. may serve, e.g., to reduce the power loss due to cross conduction of the two MOSFETs  250 ,  252 , e.g., during turn of and turn on periods. When the trigger in signal  259  is low or no trigger in signal  259  exists, the output of the IGBT gate  248  and emitter  249  may be maintained at negative rail, e.g., in order to make sure that the IGBT  46  is off and will not turn on due to electrical noise in the circuit  246 . Series gate resistors, e.g., between the MOSFETs  250 ,  252  outputs to the IGBT gate  248 , though such gate resistors (not shown) could be employed. Capacitors  290 ,  292  may be used to store energy in charging and discharging the IGBT gate  248 . 
     According to aspects of am embodiment of the present invention, as illustrated schematically and partly in block diagram form in  FIG. 5B , an improved circuit  246 ′ may be essentially the same as the circuit  246  of  FIG. 5A , with the exception that the optocoupler  258  may be replaced, e.g., with an improved optocoupler  258 ′ that may comprise an optical transmitter  260 , e.g., on the low voltage side of the circuit  246 ′ and an optical receiver  262  on the high voltage side of the circuit  246 ′ along with a comparator  264 . The optical transmitter  260  may be a model HFBR-1527, made by AGILENT TECHNOLOGIES, U.S.A. and the optical receiver  262  may be a model HFBR-2526, made by AGILENT TECHNOLOGIES U.S.A., and the comparator may be a model MAX961ESA, made by MAXIM, U.S.A. 
     Together the optical transmitter  260  and optical receiver  262 , e.g., at higher operating pulse repetition rates, e.g., at about 6 kHz and above may be employed to provide a better voltage isolation between the high voltage side and the low voltage side, because the voltage isolation can be scaled by the length of the optical fiber cable between optical transmitter and optical receiver (as compared with the optocoupler  258  in  FIG. 5A  where the voltage isolation is limited by the device capabilities). The comparator  264  may serve to condition the signal from the optical receiver to make the signal amplitude large enough for the input of the MOSFET driver  254 . 
     Turning now to  FIGS. 6 and 7  there is shown, respectively a plan view of a cooling plate  300  for a step up transformer  56  according to aspects of an embodiment of the present invention. The cold plate  300  may have a plurality of sections  302 ,  304 ,  306  and  308 , each corresponding to a section of the step up transformer, i.e.,  56   a ,  56   b ,  56   c , and  56   d , corresponding generally to the sections  407 ,  408 ,  409  and  410 , shown, e.g., in  FIG. 12 , except, e.g., for the number of pucks in each section and the angle of the high voltage final section  410 ,  56   b  with respect to the preceding section  56   c ,  409 . These sections are shown by way of example to be laid out in a loop around a step up transformer mounting board  314 , e.g., to maximize space utilization on the board  314  for the placement of the sections of the step up transformer  56  totaling a certain number of primary winding pucks. It will be understood that depending on, e.g., the number of primary winding pucks needed, the size of each, the space occupied by other circuit elements, etc. the step-up transformer may be laid out in generally an L shape, e.g., with sections  56   a , and  56   b  shown in  FIG. 7 , or a generally L shaped configuration, e.g., with the sections just mentioned and also section  56   c  or in generally a loop or O-shaped configuration adding section  56   d.    
     In operation, e.g., the first cooling channel  302   a  upstream of the cooling fluid inlet  210  would contain the coldest water circulating through the cooling fluid system and the cooling channel  302   b  the hottest water circulating through the cooling water system to the cooling fluid outlet  212 , the second coolest fluid of the incoming water stream would be in the cooling channel  304   a  and the third hottest outlet fluid would be flowing in the outlet cooling channel  304   b . Similarly the third coolest inlet cooling fluid and the second hottest outlet cooling fluid would be flowing, respectively in inlet cooling channel  306   a  and outlet cooling channel  306   b , and the fourth coolest inlet cooling water would be flowing in the inlet cooling channel  308   a , and the fourth hottest cooling fluid would be flowing in the outlet cooling channel  308   b . In this manner approximately on average each section  302 ,  304 ,  306  and  308  would have about the same capacity to transfer heat away from its adjoining transformer  56  section  56   a ,  56   b ,  56   c  and  56   d . In this arrangement also, e.g., the coolest water entering through cooling fluid input, e.g., from a coolant fluid supply conduit  320 , in the cold plate section  302  may serve to also provide some heat removal from the proximate section  308 , which, e.g., may be the coolant plate  300  section over the hottest portion  56   d  of the step-up transformer  56 . 
     It will be understood that the cooling fluid may be a liquid or a gas, though a liquid is preferred and water is used according to aspects of an embodiment of the present invention. In addition, it will also be understood that the cooling channels may be formed to make a plurality of loops around and back through the respective number of sections of the cold plate  300 , either from the same single coolant fluid inlet  310  to the same coolant fluid outlet  312  or from a plurality of such coolant inlets and outlets, on pair for each loop of inlet and outlet channels, or a combination thereof. It will also be understood that the coolant channels, e.g., cooling channel  302   a , cooling channel  302   b . cooling channel  304   a , cooling channel  304   b , cooling channel  306   a , cooling channel  306   b , cooling channel  308   a , and cooling channel  308   b  could be formed in a variety of ways, e.g., by forming a channel in at least one half of a cold plate  300 , illustrated by way of example in  FIG. 6 , and joining it with another half of the cold plate  300  (not shown), which may or may not have matching channels, e.g., by vacuum brazing, to form a very strong single piece cold plate with internally formed channels. Alternatively, by way of example, the channels may simply be formed as, e.g., grooves  332  in the surface of the cold plate  300 , e.g., with the cooling fluid flowing through the channels in a thermally conductive tubing, e.g., a copper tubing, e.g., pressed into the grooves  332 , e.g., as illustrated in  FIG. 7 . 
     It will be understood that the cold plate  300  may be attached to the transformer  56 , e.g., by extending the length of at least one side  460 ′ of the pucks  402  to meet the cold plate and attaching the cold plate  300  to such extended sides of the respective puck  402 , e.g., by a thermally conductive adhesive, such as Silver Conductive Grease made by ITW Chemtronics, U.S.A., such as is shown in FIGS.  8 ,  10  and  11  Alternatively, e.g., such adhesive could be used to connect the cold plate  300  also to the puck insulators  460 , such as are shown in  FIG. 11 . 
     According to aspects of an embodiment of the present invention, applicants have found that during high voltage operation of the SSPPM transformer  56 , corona or partial discharge can develop in the transformer  56  assembly (particularly in the region between the transformer secondary winding  400  and the individual pucks  402 ). The pucks  402  may each contain a single winding  404  (as shown for example in  FIG. 7 ) or a pair of windings  406  (as shown, e.g., in  FIGS. 10 and 11 ). Such corona discharges can be exaggerated in the pucks  402  at the high voltage end  410  of the secondary since the voltages are higher there between the secondary  400  and the respective primary, formed by the respective pucks  402 . 
     Previously applicants&#39; assignee&#39;s lasers systems have employed extruded coaxial cable  430 , shown, e.g., in  FIG. 11 , so that most of the electrical field is seen in the polyethylene insulation  420  associated with the coaxial cable piece  430 . Because the polyethylene is extruded over the cable center conductor  432 , no air gap is allowed to exist at that location where partial breakdown or corona might develop (the polyethylene  420  also has a higher breakdown strength than air). An air gap  440  does exist at the outer diameter of the cable piece  430  (the outer diameter of the polyethylene  420 ) between that and the inner diameter of the respective transformer pucks  402  and respective end flanges  470 . As the operating voltages increase, the fields must either be reduced in this location (by making the parts bigger) or else the insulation must be improved so that corona or breakdowns do not occur. Since neither of these solutions is particularly attractive in the applications noted herein, applicants, according to aspects of an embodiment of the present invention propose to enclose the entire transformer secondary section  400  (at least in the last transformer leg where the voltages are the highest) and then to also provide that region with pressurized insulation gas or with a liquid insulation filling to allow higher breakdown fields in the region. 
     A solution, as illustrated, e.g., in  FIGS. 10-16 , according to aspects of an embodiment of the present invention, can, e.g., allow either pressurized gas or liquid insulation. O-rings  450  may be added between each transformer puck  402  face  452  and an insulator  460  (shown in more detail in  FIG. 16  between each puck  402 . In addition, flanges  470  may be added to mate against the end puck  402 ′ faces  452 ′ on the ends of at least the last high voltage leg  410  in the transformer  56 . If necessary, each transformer  56  leg including the transformer legs  405 ,  407  and  409 , as shown, e.g., in  FIG. 12  (or as many as required based on the state of the voltage in each) could also be sealed and insulated in a similar manner to ensure a corona discharge did not develop in any location of the transformer secondary  400  passing through the respective leg. 
     As can be seen from  FIGS. 10-12 , the insulating spacers between pucks  402  and the space sealed by respective o-rings  450  inserted in o-ring grooves  456  formed in the puck faces  452 . Similarly, the end flanges  470 , shown in more detail in, e.g.,  FIGS. 14 and 15  may have cable o-ring seal grooves  460  into which cable o-ring seals (not shown) may be inserted to seal the ends of the respective secondary winding section  405 ,  407 ,  409  and  410 . At least one of the end flanges  470  may have an opening  480  which may form the inlet for the insulating gas, e.g., N 2  or fluid, e.g., oil, or one opening  480  on one end may an inlet for the insulating gas/liquid and the other may be the outlet port. 
     Turning now to  FIG. 17  there is shown schematically a more detailed view of a portion of the circuitry shown in  FIG. 1 , i.e., a prior art saturable assist circuit for biasing the partly saturable inductor L 0    48 , e.g., away from saturation between openings of the solid state switch  46  to transfer the energy from capacitor C 0    42  into the pulse compression circuitry, including, e.g., first onto a first stage capacitor C 1    52  and the transformer  56 . As noted above, the solid state switch may be a plurality of solid state switches, e.g., two solid state switches  42 ′ and  42 ″ in parallel, connected to charging capacitor C 0-42 . The saturable inductor L 0    48  and diode  58  may be a pair of saturable inductors  48 ′ and  48 ″ in series with the solid state switch  42 ′ and  48 ′″ and  48 ″″ in series with the solid state switch  42 ″. The diode  58  may be a pair of diodes  58 ′ and  58 ″ in series with the solid state switch  46 ′ and a pair of diodes  58 ′″ and  58 ″″ in series with the solid state switch  46 ″. Each of the charging inductors L 0    48 ′,  48 ″,  48 ′″ and  48 ″″ may in turn be made up of a first saturable inductor  48 ′ a ,  48 ″ a ,  48 ′″ a  and  48 ″″ a  and a second saturable inductor  48 ′ b ,  48 ″ b ,  48 ′″ b  and  48 ″″ b , and an inductor  48 ′ c ,  48 ″ c ,  48 ′″c and  48 ″″c. 
     Each of the diodes  58 ′,  58 ″,  58 ′″ and  58 ″″ may comprise a pair of series connected diodes  58 ′ a  and  b ,  58 ″ a  and  b ,  58 ′″ a  and  b  and  58 ″″ a  and  b , each with a respective resonance bypass circuit. Each of the charging inductors L 0    48 ′,  48 ″,  48 ′″ and  48 ″″ has in this prior art circuit  120  that may be used to bias respective ones of the inductor pairs  48 ′,  48 ″ and  48 ′″,  48 ″″, e.g., by being magnetically connected, respective to the cores of saturable inductors  48 ′ a  and  b  on the one hand and  48 ″″ a  and  b  on the other. 
     Turning now to  FIG. 18 , there is shown an improved circuit to that of  FIG. 17 , wherein, e.g., the diode arrays  58 ′ and  58 ″ have been replaced with a single diode  58 ′ and the diode arrays  58 ′″ and  58 ″″ have bee replaced with a single diode  58 ′, neither of which has a respective resonance circuit shown in  FIG. 17 . Similarly the pairs of charging inductors  48 ′ and  48 ″ and  48 ′″ and  48 ″″ have been replaced by a single charging inductor  48 ′ in series with solid state switch  46 ′ and  48 ″ in series with solid state switch  46 ″. 
     A single biasing circuit  120 ′ which may comprise a bias inductor  122 , in series with a parallel arrangement of two identical RC circuits  124  which may comprise a 24000 μF capacitor  126  across a 5V dc biasing voltage power supply  128  and series with a 0.1 ohm resistor  129 , both in parallel with a 12000 μF capacitor  130 . 
     The bias inductor  122  may also be connected in parallel with the saturable portions  48 ′ a  and  b  and  48 ″ a  and  b  of the respective charging inductors  48 ′ and  48 ″. such an arrangement, in addition to being less costly can provide for a smoother and more echonomical transition of the energy from Charging Capacitor C 0    42  to first stage capacitor C 1    52  when the solid state switches  46 ′ and  46 ″ are closed. 
     While the particular aspects of embodiment(s) of the 6K PULSE REPETITION RATE AND ABOVE GAS DISCHARGE LASER SYSTEM SOLID STATE PULSE POWER SYSTEM IMPROVEMENTS described and illustrated in this patent application in the detail required to satisfy 35 U.S.C. §112 is fully capable of attaining any above-described purposes for, problems to be solved by or any other reasons for or objects of the aspects of an embodiment(s) above described, it is to be understood by those skilled in the art that it is the presently described aspects of the described embodiment(s) of the present invention are merely exemplary, illustrative and representative of the subject matter which is broadly contemplated by the present invention. The scope of the presently described and claimed aspects of embodiments fully encompasses other embodiments which may now be or may become obvious to those skilled in the art based on the teachings of the Specification. The scope of the present 6K PULSE REPETITION RATE AND ABOVE GAS DISCHARGE LASER SYSTEM SOLID STATE PULSE POWER SYSTEM IMPROVEMENTS is solely and completely limited by only the appended claims and nothing beyond the recitations of the appended claims. Reference to an element in such claims in the singular is not intended to mean nor shall it mean in interpreting such claim element “one and only one” unless explicitly so stated, but rather “one or more”. All structural and functional equivalents to any of the elements of the above-described aspects of an embodiment(s) that are known or later come to be known to those of ordinary skill in the art are expressly incorporated herein by reference and are intended to be encompassed by the present claims. Any term used in the specification and/or in the claims and expressly given a meaning in the Specification and/or claims in the present application shall have that meaning, regardless of any dictionary or other commonly used meaning for such a term. It is not intended or necessary for a device or method discussed in the Specification as any aspect of an embodiment to address each and every problem sought to be solved by the aspects of embodiments disclosed in this application, for it to be encompassed by the present claims. No element, component, or method step in the present disclosure is intended to be dedicated to the public regardless of whether the element, component, or method step is explicitly recited in the claims. No claim element in the appended claims is to be construed under the provisions of 35 U.S.C. §112, sixth paragraph, unless the element is expressly recited using the phrase “means for” or, in the case of a method claim, the element is recited as a “step” instead of an “act”. 
     It will be understood by those skilled in the art that the aspects of embodiments of the present invention disclosed above are intended to be preferred embodiments only and not to limit the disclosure of the present invention(s) in any way and particularly not to a specific preferred embodiment alone. Many changes and modification can be made to the disclosed aspects of embodiments of the disclosed invention(s) that will be understood and appreciated by those skilled in the art. The appended claims are intended in scope and meaning to cover not only the disclosed aspects of embodiments of the present invention(s) but also such equivalents and other modifications and changes that would be apparent to those skilled in the art. In additions to changes and modifications to the disclosed and claimed aspects of embodiments of the present invention(s) noted above others could be implemented.