Patent Publication Number: US-7715295-B2

Title: Optical disc reproducing apparatus for correcting asymmetric errors in data reproduced from optical discs

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   This application claims priority from Korean Patent Application No. 10-2006-0016828, filed on Feb. 21, 2006, in the Korean Intellectual Property Office, the entire disclosure of which is incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to an optical disc reproducing apparatus, which detects binary data from analog signals obtained from an optical disc under various kinds of optical disc environments including a high-density disc environment such as 15 GB in a 12 cm diameter disc by supporting Run Length Limited (RLL)(1,10) and RLL(2,10) codes, a 5-tap partial response maximum likelihood (PRML) sixteen levels, an asymmetric compensating operation under such RLL code environments, and a minimum T signal compensating operation. 
   2. Description of the Related Art 
   With the onset of the multimedia era, the need for storing and transmitting a large quantity of digital data has increased. Accordingly, an optical disc such as a digital video disc (DVD) has been widely studied in the art. The current DVD market is steadily growing, being divided into a computer industry that desires to adopt a DVD-ROM and a home appliance industry that intends to promote a DVD-video. Additionally, the DVD extends an applicable sphere as a DVD-R (recordable), a DVD-RW (rewritable), and a DVD-RAM (random access memory) appear on the market. 
   Such kinds of conventional optical discs may often be confronted with various problems as follows. When data stored in the optical disc are reproduced, analog radio frequency (RF) signals under reproduction may frequently exhibit an asymmetric waveform. Furthermore, this asymmetric phenomenon may give rise to other unfavorable phenomena such as jitter, non-linear bit transition, DC transition, and inter-symbol interference (ISI) between symbols of reproduced data. Such phenomena may make it difficult to execute the detection and correction of frequency errors and phase errors, thus causing the distortion of reproduced signals. A conventional optical disc reproducing apparatus has typically used a digital sum value (DSV) algorithm to correct such asymmetric errors. 
   However, the conventional DSV algorithm does not always execute an exact correction of the asymmetric errors under various code environments such as RLL(1,10) and RLL(2,10). For example, the conventional DSV algorithm may often fail to exactly detect asymmetric errors in the case of 4T sampling signals reproduced in the variable frequency oscillator (VFO) sector of the optical disc. 
   Furthermore, the conventional optical disc reproducing apparatus does not support an integrated solution for both RLL(1,10) and RLL(2,10) codes. Therefore, two separate and independent code detectors, namely, adding an RLL(1,10) detector to an existing RLL(2,10) detector, are used to support the integrated solution. This configuration of the detectors may, however, be inefficient in circuit area use and power consumption. 
   Additionally, a 4-Tap partial response maximum likelihood (PRML) structure of the conventional optical disc reproducing apparatus may fail to execute the optical disc reproduction under a high-density environment of more than 15 GB in a 12 cm diameter disc. 
   Additionally, a frequency detector of the conventional optical disc reproducing apparatus may fail to effectively detect frequency errors under such a high-density environment with a frequent ISI and a high noise. Also, a delay in frequency locking timing may interrupt a stable phase locked loop (PLL) operation. 
   Additionally, an asymmetry compensator of the conventional optical disc reproducing apparatus may be unavailable for some optical discs such as DVD-RAM, thus putting restrictions on reproducible kinds of the optical disc. 
   Additionally, the above-mentioned detector of the conventional optical disc reproducing apparatus requires a high-priced external measurer to measure a signal quality, therefore, actual measurement of the signal quality is not simple. 
   Additionally, a minimum T compensator of the conventional optical disc reproducing apparatus may be not operable under an RLL(1,10) code environment. 
   To overcome the above-discussed problems, an improved optical disc reproducing apparatus capable of operating under various disc environments and capable of executing an exact and effective reproduction under a high-density environment with a frequent ISI and a high noise is required in the art. 
   SUMMARY OF THE INVENTION 
   One aspect of the present invention is to provide an optical disc reproducing apparatus capable of operating by supporting both an RLL(1,10) code and an RLL(2,10) code. 
   Another aspect of the present invention is to provide an optical disc reproducing apparatus capable of executing an effective disc reproduction under a high-density environment of more than 15 GB in a 12 cm diameter disc by supporting a 5-Tap PRML structure. 
   Still another aspect of the present invention is to provide an optical disc reproducing apparatus capable of effectively detecting frequency errors under a high-density environment with a frequent ISI and a high noise. 
   Still another aspect of the present invention is to provide an optical disc reproducing apparatus capable of executing an asymmetric compensating operation in response to an RLL(1,10) environment. 
   Still another aspect of the present invention is to provide an optical disc reproducing apparatus capable of executing a minimum T signal compensating operation under 3T and 2T environments in response to an RLL(1,0) code. 
   In order to achieve the above and other aspects, one exemplary, non-limiting embodiment of the present invention provides an optical disc reproducing apparatus, which comprises: an A/D converter which converts analog RF signals obtained from an optical disc to digital signals; an asymmetry compensator which detects 4T sampling signals, a polarity of which changes in a four-times cycle than a sampling cycle, among the digital signals, and compensates a level of the digital signals according to counted asymmetric error values; a phase locked loop which includes a frequency detector that counts and detects run-length signals from the digital signals and compensates frequency errors of the digital signals; a binary module which includes a Viterbi decoder that detects binary data from the digital signals, a slicer that determines the binary data depending on a predetermined threshold value, and a minimum T compensator that compensates the digital signal with a minimum signal having a unit cycle; an equalizer which equalizes a specific frequency of the digital signal; an adaptive level error detector which detects a base level of the Viterbi decoder from both an input signal into the equalizer and an output signal from the Viterbi decoder, and computes a filtering coefficient of the equalizer from the base level; and a signal quality measurer which measures a jitter or a simulated bit error rate (SbER) of the digital signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a view that illustrates an optical disc reproducing apparatus according to an exemplary embodiment of the present invention. 
       FIG. 2  is a block diagram that illustrates an asymmetry compensator according to an exemplary embodiment of the present invention. 
       FIG. 3  is a block diagram that illustrates a signal detector of the asymmetry compensator according to an exemplary embodiment of the present invention. 
       FIG. 4  is a view that illustrates a 4T sampling signal according to an exemplary embodiment of the present invention. 
       FIG. 5  is a view that illustrates signal waveforms used in the signal detector of the asymmetry compensator according to an exemplary embodiment of the present invention. 
       FIG. 6  is a view that illustrates a DSV method for an RLL(1,10) code according to an exemplary embodiment of the present invention. 
       FIG. 7  is a block diagram that illustrates a frequency detector according to an exemplary embodiment of the present invention. 
       FIG. 8  is a view that illustrates a run-length distribution density to a channel coding feature of the optical disc reproducing apparatus according to an exemplary embodiment of the present invention. 
       FIG. 9  is a view that illustrates run-length boundaries in the run-length distribution density. 
       FIG. 10  is a view that illustrates a shift of a run-length signal distribution in the run-length distribution density when the frequency of a sampling clock is lower than a target frequency. 
       FIG. 11  is a view that illustrates a relation among a real signal count, a predicted signal count, and a threshold value when the run-length signal distribution is shifted as shown in  FIG. 10 . 
       FIG. 12  is a view that illustrates a shift of a run-length signal distribution in the run-length distribution density when the frequency of a sampling clock is higher than a target frequency. 
       FIG. 13  is a view that illustrates a relation among a real signal count, a predicted signal count, and a threshold value when the run-length signal distribution is shifted as shown in  FIG. 12 . 
       FIG. 14  is a block diagram that illustrates a channel identifier of an adaptive level error detector according to an exemplary embodiment of the present invention. 
       FIG. 15  is a view that illustrates a trellis diagram of a 5-tap Viterbi decoder of a (1,7) code according to an exemplary embodiment of the present invention. 
       FIG. 16  is a view that illustrates a level estimation result by the Viterbi decoder in  FIG. 15 . 
       FIG. 17  is a block diagram that illustrates an SbER controller of a signal quality measurer according to an exemplary embodiment of the present invention. 
       FIG. 18  is a view that illustrates a reference table according to an exemplary embodiment of the present invention. 
       FIG. 19  is a view that illustrates a jitter controller according to an exemplary embodiment of the present invention. 
       FIG. 20  is a view that illustrates a jitter detector of the jitter controller according to an exemplary embodiment of the present invention. 
       FIG. 21  is a view that illustrates a calculation method of a jitter value according to an exemplary embodiment of the present invention. 
       FIG. 22  is a view that illustrates a cycle examiner of the jitter controller according to an exemplary embodiment of the present invention. 
       FIG. 23  is a view that illustrates a timing diagram of a counter of the cycle examiner according to an exemplary embodiment of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Reference will now be made in detail to exemplary embodiments of the present invention, examples of which are illustrated in the accompanying drawings, wherein like reference numerals refer to the like elements throughout. The exemplary embodiments are described below in order to explain the present invention by referring to the figures. 
     FIG. 1  is a view that illustrates an optical disc reproducing apparatus according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 1 , the optical disc reproducing apparatus of the current embodiment includes an A/D converter  110 , an asymmetry compensator  120 , a phase locked loop (PLL)  130 , a binary module  140 , an equalizer  150 , an adaptive level error detector  160 , and a signal quality measurer  170 . The binary module  140  includes a Viterbi decoder  141 , a slicer  142 , and a minimum T compensator  143 . The signal quality measurer  170  includes a jitter controller  171  and an SbER controller  172 . 
   The A/D converter  110  is a device that converts analog RF signals, extracted from an optical disc, to digital signals by sampling the analog RF signals in a predetermined sampling cycle. 
   The asymmetry compensator  120  detects 4T sampling signals, a polarity of which changes in a four-times cycle than the sampling cycle, among the digital signals. The asymmetry compensator  120  also calculates asymmetric error values from the 4T sampling signals and counts them. Then the asymmetry compensator  120  compensates a level of the digital signals according to the counted asymmetric error values. 
   The asymmetry compensator  120  may use a digital sum value (DSV) algorithm for the asymmetric error compensation. The DSV algorithm is a procedure for computing DSVs and then deciding whether they are asymmetric errors or not. The DSV is a value that indicates how much direct current components are contained in the digital signals. 
   To explain the DSV algorithm in detail, an optical recording/reproducing system converts the analog RF signals obtained in the optical disc into the digital signals. Then the system computes the average of two digital signals detected in sequence. If the average is positive, a polarity is assigned to “1”. And a polarity of a negative average is assigned to “−1.” Such polarities are counted and accumulated. If the accumulated value of the polarity exceeds a pre-established threshold value, this is regarded as the occurrence of the asymmetric error. Finally, input signals are compensated. 
   This technique of compensating the asymmetric errors by using the DSV algorithm has been disclosed in the Korean Patent Publication No. 2001-0035777. The DSV algorithm in the above disclosure may fail to execute the asymmetric error compensation under the RLL(1,10) environment. The asymmetry compensator  120  of the present invention is capable of executing such asymmetric error compensation in response to the RLL(1,10) environment. Related explanations will follow referring to  FIGS. 2 to 6 . 
     FIG. 2  is a block diagram that illustrates an asymmetry compensator according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 2 , the asymmetry compensator  200  of an exemplary embodiment of the invention includes a signal detector  210 , a calculator  220 , an error size adjuster  230 , a counter  240 , and a corrector  250 . 
   The signal detector  210  extracts the 4T sampling signals from the digital signals that the A/D converter outputs. The 4T sampling signal means a digital signal in a four times sampling region every half cycle of the analog RF signal. That is, when the sampling cycle of the A/D converter is indicated by “T”, a digital signal that changes its polarity in a 4T cycle may become the 4T sampling signal. 
   The calculator  220  computes the asymmetric error values by adding selected ones among the 4T sampling signals. Because of the 4T sampling signals, eight digital signals can be detected from every cycle of the analog RF signal. The calculator  220  may add first, fourth, fifth, and eighth digital signals among eight digital signals and then compute the asymmetric error value. Unless the asymmetric error happens, the asymmetric error value is computed to “0”. If the asymmetric error happens, the asymmetric error value is computed to a predetermined value except “0.” 
   To compute the asymmetric error value, the calculator  220  may add all of eight digital signals, not first, fourth, fifth, and eighth only. Furthermore, the calculator  220  may compute the asymmetric error value by adding two symmetric digital signals on opposite sides of a point where the size of the analog RF signal is zero. 
   The error size adjuster  230  regulates the size of the asymmetric error value by multiplying a certain adjustment coefficient. The magnitude of the adjustment coefficient may be predetermined. If the adjustment coefficient is determined to be “1”, the asymmetric error value is transmitted to the counter  240  as it is. If the adjustment coefficient is determined to be “0.5”, the asymmetric error value is reduced in half and transmitted to the counter  240 . Depending on the applicable spheres of the optical disc reproducing apparatus of the invention, the magnitude of the adjustment coefficient may be arbitrarily determined by those skilled in the art. 
   The counter  240  counts the asymmetric error values by accumulating them as they are outputted from the error size adjuster  230 . 
   Unless the asymmetric error values fall within a predetermined critical range, the corrector  250  adjusts a signal level of the digital signals outputted from the A/D converter by the asymmetric error values. The asymmetric errors are therefore compensated. When executing the asymmetric error compensation, the corrector  250  may renew the asymmetric error value that the counter  240  has counted. 
     FIG. 3  is a block diagram that illustrates a signal detector of the asymmetry compensator according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 3 , the signal detector of an exemplary embodiment of the invention includes a variable frequency oscillator (VFO) detecting unit  310  and a VFO sampling signal output unit  320 . The VFO detecting unit  310  has a first header signal detecting section  311 , a second header signal detecting section  312 , a pulse output section  313 , a timer section  314 , and a VFO detection signal generating section  315 . 
   The VFO detecting unit  310  outputs VFO detection signals while a variable frequency oscillator (VFO) sector of the optical disc is reproduced. The VFO sector stores VFO signal data in the sectors of the optical disc. The VFO signal allows the determination of whether a region of the optical disc is a land or a groove. Namely, a negative VFO signal is determined to be a land region, and a positive VFO signal is determined to be a groove region. Also, the VFO signal becomes a base signal used for generating exact synchronous signals in the phase locked loop ( 130  in  FIG. 1 ). For this, the VFO signal changes its polarity in a 4T cycle. 
   The VFO sector is located at a header field and a user data field in the optical disc, such as the DVD-RAM. The header field is classified into a header peak area and a header bottom area. Such header areas allow knowing whether the next track is the land or the groove. Namely, if the next track is the land, a header peak signal is generated, and if the next track is the groove, a header bottom signal is generated. The VFO sector is located at respective front portions of the header peak area, the header bottom area, and the user data field. 
   The first and second header signal detecting sections  311  and  312  detect respectively the header peak signal reproduced at the header peak area and the header bottom signal reproduced at the header bottom area. 
   The pulse output section  313  outputs a control pulse when at least one of the header peak signal and the header bottom signal is detected in at least one of the first and second header signal detecting sections  311  and  312 . 
   The timer section  314  begins counting actions when the control signal is inputted from the pulse output section  313 . Then the timer section  314  counts an elapsed time from a point when the header peak signal or the header bottom signal is detected. 
   If a time counted by the timer section  314  exceeds a first time, the VFO detection signal generating section  315  outputs the VFO detection signal for a predetermined time. Furthermore, if a second and a third times are pre-established to the respective VFO sectors located at the header bottom area and the user data field, the VFO detection signal generating section  315  can output the VFO detection signal at each reproducing time. 
   The VFO sampling signal output unit  320  outputs the digital signals, outputted from the A/D converter, as the 4T sampling signals while the VFO detection signal is inputted from the VFO detecting unit  310 . 
     FIG. 4  is a view that illustrates a 4T sampling signal according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 4 , eight sampling actions are carried out in a single cycle of the analog RF signal. Therefore, eight digital signals, i.e., D 0  to D 7 , are detected from the A/D converter. Among them, the calculator  220  adds D 0 , D 3 , D 4 , and D 7 . The result of adding becomes the asymmetric error value. The calculator  220  may add all of D 0  to D 7 , and also may add two symmetric signals, such as D 3  and D 4 , on opposite sides of the zero point. 
     FIG. 5  is a view that illustrates signal waveforms used in the signal detector of the asymmetry compensator according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 5 , the header peak signal is detected for a time interval between t 0  and t 4 , and the header bottom signal is detected for another time interval between t 3  and t 7 . Therefore, the pulse output section  313  outputs the control pulse for a time interval between t 0  and t 7 . 
   The VFO detection signal generating section  315  outputs the VFO detection signal at a first time point t 1  after a predetermined time elapses from an initial time point t 0  when the control pulse is inputted. The first time point t 1  is determined by considering a data reproducing and transmitting time, a converting time, and so forth. Since the output time of the VFO detection signal is delayed by a time difference between t 1  and t 0 , an exact detection of the 4T sampling signal is possible. 
   Since no 4T sampling signals are detected after the VFO sector is reproduced, the conventional DSV algorithm may be favorably used to compensate the asymmetric errors. For this, the VFO detection signal generating section  315  outputs the VFO detection signal only for a time difference between t 2  and t 1 , corresponding to a length of the VFO sector. Consequently, , the asymmetric error values are computed, accumulated, and counted only while the VFO detection signal is outputted, and the asymmetric error compensating operation is executed to compensate signal levels by using counting results. 
   Additionally, since the VFO sector exists in the header bottom area, the VFO detection signal is outputted for a time difference between t 6  and t 5  from t 5  corresponding to the header bottom area. That is, when a time counted from t 0  by the timer section  314  is equal to a difference between t 5  and t 0 , the VFO detection signal generating section  315  outputs again the VFO detection signal. Also, when a counting time is equal to a difference between t 6  and t 0 , the output of the VFO detection signal is stopped. 
   Since the VFO sector further exists in the user data field, the VFO detection signal is outputted again after a time elapses by a difference between t 8  and t 0 . Also, the output of the VFO detection signal is stopped again after a time elapses by a difference between t 9  and t 0 . 
   A signal level compensating method by the asymmetric error detection of the aforementioned asymmetry compensator is fully described referring to  FIG. 6 . 
     FIG. 6  is a view that illustrates a DSV method for an RLL(1,10) code according to an exemplary embodiment of the present invention. 
   As shown in  FIG. 6 , the asymmetry compensator ( 120  in  FIG. 1 ) may include a data detector  610 , a high gain detector  620 , a 4T detector  630 , and a subtracter  640  by a functional classification. 
   The data detector  610  receives the digital signal converted from the analog RF signal. The high gain detector  620  detects high gains from the header peak and the header bottom. The 4T detector  630  detects the 4T sampling signal from the digital signal. 
   The 4T detector  630  creates a flag informing the detection of the 4T sampling signal in case where the high gain is “1”. Whenever the flag is created, the 4T detector  630  adds four sample values of the 4T sampling signal and thus creates an asymmetric tracking value. 
   Since a single asymmetric tracking value is insufficient for satisfying accuracy, the 4T detector  630  may create a final asymmetric tracking value by multiplying a gain and the asymmetric tracking value. 
   The subtracter  640  subtracts the final asymmetric tracking value from the asymmetric level when the flag is created. However, if the flag is not created, the subtraction of the asymmetric tracking value is not carried out. 
   Thus the optical disc reproducing apparatus of the present invention can solve asymmetric tracking issues in the header sector and in the user data field under the RLL(2,10) environment by the operation of the asymmetry compensator. 
   The configuration and the operation of the asymmetric compensator of the present invention may be embodied to involve those of “apparatus for correcting defect level and asymmetric waveform level in optical disk system” disclosed in the Korean Patent Publication No. 2001-0035777. 
   The phase locked loop  130  shown in  FIG. 1  includes a frequency detector, which counts and detects run-length signals from the sampling digital signals during a frequency detection cycle according to a run-length distribution density depending on a channel coding feature. Also, the frequency detector generates frequency errors during the frequency detection cycle through the count value of the run-length signals and other referential values, and compensates the frequency errors of the digital signals. 
   The configuration and the operation of the frequency detector are fully described hereinafter referring to  FIGS. 7 to 13 . 
     FIG. 7  is a block diagram that illustrates a frequency detector according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 7 , the frequency detector includes an edge counter  710 , a run-length signal detection unit  720 , a counter unit  730 , and a frequency error generation unit  740 . 
   The run-length signal detection unit  720  receives the sampled RF signal from the A/D converter. The sampled RF signal means a digital signal sampled from the analog RF signal by the A/D converter, as discussed above. In the following descriptions referring to  FIGS. 7 to 13 , such a digital signal will be referred to as the sampled RF signal. 
   When the sampled RF signal is inputted, the edge counter  710  controls the frequency detection cycle by counting passing edges of the sampled RF signal. The passing edges include a rising edge and a falling edge. When a value counting the passing edges of the sampled RF signal reaches a predefined number, the edge counter  710  controls the frequency detection cycle by outputting a positive pulse. 
   After the positive pulse is outputted, the edge counter  710  clears the edge counting value and counts again the passing edges of the sampled RF signal to control the next frequency detection cycle. Unless the counting value reaches the predefined number, the output of the edge counter  710  always maintains “0”. The positive pulse is transmitted to the counter unit  730  and the frequency error generation unit  740 . 
   If the predefined number is high, an accuracy of evaluation is improved but a frequency error detection speed is reduced. If the predefined number is low, the accuracy is reduced but the speed is improved. Therefore, the predefined number may be determined on the basis of a range for evaluating the run-length distribution density with regard to the sampled RF signal. 
   The run-length signal detection unit  720  detects a run-length signal from the sampled RF signal based on the run-length distribution density predicted according to the channel coding feature of the optical disc reproducing system. Specifically, the run-length signal detection unit  720  divides a run-length region into at least two partitions, based on the predicted run-length distribution density. Then the run-length signal detection unit  720  detects the run-length signal of the RF signal, sampled during the frequency detection cycle, according to the run-length partitions. 
   In the run-length signal detection unit  720  exemplarily shown in  FIG. 7 , the run-length region is divided into (n-1) pieces of partitions from 2T to nT and the other partition. 
   In case of the run-length distribution density shown in  FIG. 8 , the run-length signal detection unit  720  may divide the run-length region into n pieces of partitions by using predefined run-length boundaries (2T_up, 3T_up, . . . , nT_up) as shown in  FIG. 9 . 
   The number “n” may be determined on the basis of the maximum run-length of various types of the optical discs employed in the apparatus of the invention. For example, if the apparatus of the invention uses a CD, a DVD, a BD (blue ray), and a HD-DVD, the maximum run-length of which is 11T, 11T, 8T, and 11T, respectively, the number “n” is determined to “11”. 
   Alternatively, the run-length signal detection unit  720  may divide the run-length region into the minimum run-length partition (e.g., 2T) and the other run-length partitions so as to detect the run-length signal. 
   Furthermore, the run-length signal detection unit  720  may divide the run-length region into the most frequently generating run-length partition and the other run-length partitions. In  FIGS. 8 and 9 , the 2T run-length partition becomes the most frequently generating run-length partition, however any other run-length partition may alternatively become the most frequently generating run-length partition. 
   The counter unit  730  includes at least one counter that counts the run-length signal detected in the run-length signal detection unit  720  during the frequency detection cycle. In a  FIG. 7  example, the counter unit  730  has n pieces of counters ( 730 _ 1 ˜ 730 _n) because the number of the run-length partitions used in the run-length signal detection unit  720  is “n”. The number of the counters contained in the counter unit  730  depends on the number of the run-length partitions used in the run-length signal detection unit  720 . 
   Accordingly, in the  FIG. 7  example, since the run-length signal detection unit  720  uses (n-1) pieces of the partitions from 2T to nT and the other partition, the n pieces of the counters  730  includes a 2T counter  730 _ 1  corresponding to the 2T run-length, a 3T counter  730 _ 2  corresponding to the 3T run-length, an nT counter  730 _n-1 corresponding to the nT run-length, and an other counter  730 _n corresponding to the other run-length. 
   The 2T counter  730 _ 1  counts up whenever the 2T signal outputted from the run-length signal detection unit  720  is a positive pulse. The 3T counter  730 _ 2  counts up whenever the 3T signal outputted from the run-length signal detection unit  720  is a positive pulse. The nT counter  730 _n-1′ counts up whenever the nT signal outputted from the run-length signal detection unit  720  is a positive pulse. The other counter  730 _n counts up whenever the other signal outputted from the run-length signal detection unit  720  is a positive pulse. 
   Such counting values from the 2T counter  730 _ 1  to the other counter  730 _n are cleared when the edge counter  710  provides a positive pulse thereto. Accordingly, the value counted by the 2T counter  730 _ 1  indicates how many times the 2T run-length signal occurs during the frequency detection cycle. Similarly, the values counted by the 3T counter  730 _ 2 , the nT counter  730 _n-1, and the other counter  730 _n indicate respectively how many times the 3T run-length signal, the nT run-length signal, and the other run-length signal occur respectively during the frequency detection cycle. 
   If there are two run-length partitions that the run-length signal detection unit  720  uses, the counter unit  730  also has two counters corresponding to the run-length partitions to implement the above-discussed operation. 
   The frequency error generation unit  740  generates frequency errors during the frequency detection cycle by using predetermined threshold values (2T_thr˜other_thr) as well as the aforesaid values counted by n pieces of the counters  730 _ 1 ˜ 730 _n. 
   The threshold values (2T_thr˜other_thr) are critical points determined on the basis of the predicted run-length distribution density that may appear in the run-length region during the frequency detection cycle. That is, if the frequency of the sampling clock outputted from the phase locked loop ( 130  shown in  FIG. 1 ) is lower than a target frequency, the run-length distribution density in each run-length partition separated by the predefined run-length boundaries (2T_up, 3T_up, . . . , nT_up) as shown in  FIG. 9  may shift leftward as shown with a dotted line in  FIG. 10 . 
   As shown in  FIG. 11 , in this shifted run-length distribution density, except a real count 2T_count of the run-length signals occurring in the 2T run-length partition, remaining real counts (3T_count˜other_count) of the run-length signals occurring in another run-length partitions may be decreased in comparison with predicted counts (3T_ideal˜other_ideal). 
   Therefore, a detection of the frequency errors uses the threshold values (2T_thr˜other_thr) higher than the predicted run-length signal counts.  FIG. 11  is a view that illustrates a relation among a real signal count (nT_count) detected in each run-length partition, a predicted signal count (nT_ideal), and a threshold value (NT_thr) determined on the basis of the predicted signal count (nT_ideal), when the run-length signal distribution is shifted as shown in  FIG. 10 . 
   On the other hand, if the frequency of the sampling clock outputted from the phase locked loop ( 130  shown in  FIG. 1 ) is higher than the target frequency, the run-length distribution density in each run-length partition may shift rightward as shown with a dotted line in  FIG. 12 . When the run-length signal distribution is shifted as shown in  FIG. 12 , a relation among a real signal count detected in each run-length partition, a predicted signal count, and a threshold value determined according to the predicted signal count is illustrated in  FIG. 13 . 
   As seen from  FIG. 13 , in a run-length partition corresponding to a relatively shorter run-length signal such as 2T, a count of actually occurring run-length signals may be decreased in comparison with the predicted count. Furthermore, in another run-length partition corresponding to a longer run-length signal than 2T, a count of actually occurring run-length signals may be increased in comparison with the threshold value determined on the basis of the predicted count. 
   If the run-length count and the threshold value sampling clock frequency obtained from the corresponding partition are higher than the target frequency, the count of the run-length partition corresponding to a run-length signal longer than 2T may be higher than the threshold value. Also, the count of the 2T partition and the count of the run-length partition corresponding to a run-length signal shorter than 2T may be lower than the threshold value. When the frequency of the sampling clock outputted in the PLL circuit is increased, the counts of the run-length partitions corresponding to shorter and shorter run-length signals may be lower than the corresponding threshold value in each partition. 
   Considering the threshold value and the count of actually detected run-length signals, the threshold value can be determined. 
   Returning to  FIG. 1 , the binary module  140  includes the Viterbi decoder  141 , the slicer  142 , and the minimum T compensator  143 . 
   The Viterbi decoder  141  detects binary data from the digital signals. A partial response (PR) type of the Viterbi decoder  141  may be established to PR (a, b, c, d, e). 
   The slicer  142  detects the binary data based on a predetermined threshold value. To detect the binary data from the digital signal, the slicer  142  computes a slicing level signal by integrating an average of the digital signals, and the Viterbi decoder  141  compares the slicing level signal with the digital signal. Furthermore, the slicer  142  may detect the binary data directly from the output of the equalizer  150  to the slicing level signal. The Viterbi decoder  141  may also detect the binary data from the output of the equalizer  150 . 
   That is, each component of the binary module  140  may become an independent data detector capable of detecting the binary data by itself. The speed of detecting the binary data from an input in the binary module is faster in order of the slicer  142 , the minimum T compensator  143 , and the Viterbi decoder  141 . However, the efficiency of detecting the binary data follows the inverse order to the above. 
   The aforesaid binary module  140  may be embodied to involve a binary operation of a Viterbi decoder disclosed in the Korean Patent Publication No. 2005-0026320 entitled “Device and method for data reproduction”. 
   When the digital signal has a smaller cycle than a unit cycle of the minimum signal corresponding to a code of the optical disc, the minimum T compensator  143  compensates the digital signal with the minimum signal having the unit cycle. 
   Specifically, if the unit cycle T of the minimum signal is 2T, the minimum T compensator  143  may remove a 1T digital signal by controlling a path of the 1T digital signal through a switch. Furthermore, if the unit cycle T of the minimum signal is 3T, the minimum T compensator  143  may remove both a 1T digital signal and a 2T digital signal by controlling their paths through the switch. 
   The minimum T compensator  143  may be embodied to involve the configuration and the operation of a conventional minimum T compensator disclosed in the Korean Patent Publication No. 2004-10090 entitled “Apparatus and method for detecting binary data.” 
   The equalizer  150  equalizes a specific frequency of the digital signal. 
   The adaptive level error detector  160  detects a base level of the Viterbi decoder  141  from both an input signal into the equalizer  150  and an output signal from the Viterbi decoder  141 . Additionally, the adaptive level error detector  160  computes a filtering coefficient of the equalizer  150  from the base level, the input signal of the equalizer, and the output signal of the equalizer. 
   The adaptive level error detector  160  may include a channel identifier and an adaptive processor. The channel identifier detects the base level of the Viterbi decoder  141  from both the input signal into the equalizer  150  and the output signal from the Viterbi decoder  141 . The adaptive processor computes the filtering coefficient of the equalizer  150  from the base level, the input signal of the equalizer, and the output signal of the equalizer. 
   Hereinafter a configuration and an operation of the adaptive level error detector  160  are explained in detail referring to  FIGS. 14 to 16 . 
     FIG. 14  is a block diagram that illustrates a channel identifier of an adaptive level error detector according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 14  together with  FIG. 1 , the channel identifier includes a selection signal generator  1430 , a level selector  1450 , and average filters  1440 . The selection signal generator  1430  receives the output signals of the Viterbi decoder  141  and then generates a selection signal  1431 . Here, the output signal of the Viterbi decoder  141  is a binary signal having “0” or “1” and the final output decoded by the Viterbi decoder  141 . By the operation principle of the Viterbi decoder  141 , the output signal of the Viterbi decoder  141  has a connection with the input signal thereof, namely, the output signal of the equalizer  150 . This means that the output signal of the Viterbi decoder  141  may specify the kind of levels inputted into the Viterbi decoder  141 . Therefore, the input of an adaptive level may be selectively used in the binary module  140  including the Viterbi decoder  141 . 
   For example, when a signal level is generated by a PR(1,2,1) and a code type is (1,7), possible ideal level values are 4, 2, −2, and −4. If input signal levels are 4, 4, 4, 2, −2, −4, −4, −4, −2, 2, . . ., the output signals of the Viterbi decoder  141  may be 1, 1, 1, −1, −1, −1, −1, −1, 1, 1, . . . . Then, by performing a number of multiplexing operations equal to the number of the taps of the Viterbi decoder  141 , the output signals of the Viterbi decoder  141  are 111, 11-1, 1-1-1, -1-1-1, . . . , which equate to 111, 110, 100, 000, in binary signals. 
   Thus these binary signals mean that 4, 2, −2, −4, . . . are inputted respectively. After all, 111, 110, 100, 000, . . . may be used as the selection signal specifying the kind of the input levels such as 4, 2, −2, −4, . . . . 
   The output signal of the Viterbi decoder  141 , inputted into the channel identifier, is delayed and split by delays  1460 , the number of which is smaller by one than that of taps of the Viterbi decoder, and then inputted into the selection signal generator  1430 . The delayed input signals  1421 ,  1422 , . . . are merged again by the selection signal generator  1430  and generate the selection signal  1431  in the form of the binary signal. For example, if the number of the taps of the Viterbi decoder  141  is three and if the number of the delays is two, the selection signal  1431  assumes the form of 111, 110, 100, 000, . . . . The output of the Viterbi decoder  141  is not produced immediately at a time, but produced after a computation time corresponding to a defined system clock elapses. The delays  1460  are therefore used because a delay time corresponding to the computation time should be allotted to the input signal of the channel identifier in order to select the input signal corresponding to the output signal of the Viterbi decoder  141 . 
   Additionally, the selection signal  1431  may be removed in a case where it conforms to a removable Viterbi path according to conditions of the minimum signal. For example, in case of a three-tap structured Viterbi decoder using a (1,7) code, two selection signals 010 and 101 corresponding to 1T are removed, and six selection signals 000, 001, 011, 100, 110, and 111 are available. 
   Similarly, a five-tap structured Viterbi decoder using the (1,7) code needs only sixteen levels, and also sixteen selection signals are produced. As for an output of a normal Viterbi decoder  141 , an additional separate configuration is not required for a generation of the selection signal  1431  since the output signal itself of the Viterbi decoder  141  is not produced in the form of 1T. 
   Another input signal of the channel identifier is an input signal of the equalizer  150 . The input signal of the equalizer  150  has continuous values and is the target of decoding. This input signal has a real value that is different from the ideal base level. This input signal of the channel identifier passes through delays  1411 ,  1412 , the number of which corresponds to the number of memories of the Viterbi decoder  141  and is then inputted into the level selector  1450 . Based on the selection signal  1431 , the level selector  1450  transmits the input signal to the respective average filters  1440 . Each average filter  1440  corresponds to each level of the Viterbi decoder  141 . Therefore, the number of the average filters  1440  may be equal to that of the levels of the Viterbi decoder  141 . In this case as well, unnecessary paths may be removed. 
   The average filter  1440  calculates the average of the selected input signal  1441 ,  1442 ,  1443 , for a given period and then produces the calculated average as a new level output  1451 ,  1452 ,  1453 . Generally the average filter  1440  may use a low pass filter and its property of estimating a DC average value. 
   An alternative average filter  1440  may be embodied by means of the following equation 1.
 
 L′=L+ ( I−L )/ C   [Equation 1]
 
   Here, L′ means a level renewed by a newly entered input signal, and L means a former level. In addition, I and C mean a delayed input signal and a constant, respectively. As a constant C is increased, a renewing level L′ varies very slightly and therefore a degree of tracking becomes reduced. 
   The detected new levels  1451 ,  1452 , . . . shown in  FIG. 14  are inputted into the adaptive processor, which creates a new coefficient of the equalizer  150 , based on the level error to be detected. The level error to be detected means a difference between the output signal of the equalizer and a detected level. The new coefficient of the equalizer may be computed by renewing a former coefficient through a least mean square (LMS) method. This is represented in the following equation 2.
 
 Wk+ 1= Wk+ 2 *μek*Xk   [Equation 2]
 
   Here, ‘Wk+1’, ‘Wk’, ‘μ’, ‘ek’, and ‘Xk’ mean the new coefficient of the equalizer, the former coefficient of the equalizer, a tracking velocity (a real number), a difference between the output signal of the equalizer and the detected level, and the input signal of the equalizer, respectively. 
   The input signal Xk of the equalizer  150  is inputted into the adaptive processor after being delayed by the delays because a time delay happens while the adaptive processor detects a new level. 
   The tracking velocity μ is a parameter that determines a degree of tracking, and may be adjustable by a suitable controller. As the tracking velocity μ is higher, a degree of tracking becomes increased. 
   The adaptive processor forces a channel to be stable. The channel identifier generates an optimum level of the Viterbi decoder  141  on the basis of the input signal of the equalizer  150 . Furthermore, the adaptive processor adjusts again the coefficient of the filter by using the optimum level and thereby eliminates noises only while forcing the output signal of the equalizer  150  to nearly maintain the frequency feature of the original channel. This may allow offering much higher stability to a stabilization of coefficients or a divergence issue of the LMS algorithm that has caused problems. 
     FIG. 15  is a view that illustrates a trellis diagram of a 5-tap Viterbi decoder of a (1,7) code according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 15 , it is appreciated that paths of the 1T signal are removed. Therefore, the total number of the paths is sixteen, and further, the number of the levels is also sixteen. 
     FIG. 16  is a view that illustrates a level estimation result by the Viterbi decoder in  FIG. 15 . 
     FIG. 16  shows sixteen ideal levels 00000, 00001, 00011, 00110, 00111,. The signals entered into the channel identifier are 39, 37, −18, −68, . . . , and here the selection signals are 11100, 11000, 10000, 00000, 00001,. The number of the selection signals is equal to that of the levels. By selecting the levels being currently computed according to the selection signals, the selected level signals come to 47 (in case of 11100), 27 (in case of 11000), −22 (in case of 10000), −63 (in case of 00000). 
   Namely, the selected level signals are quite similar with the input signals. Also, the most ideal level value may be obtained by calculating the average of the delayed input signals of the channel identifier by means of the above equation 1. 
   The aforesaid adaptive level error detector  160  may be embodied to involve the configuration and the operation of a channel identifier and an adaptive processor disclosed in the Korean Patent Publication No. 2005-0026320 entitled “Device and method for data reproduction”. 
   Returning to  FIG. 1 , the signal quality measurer  170  includes the jitter controller  171  and the simulated bit error rate (SbER) controller  172 . The signal quality measurer  170  may measure a jitter or an SbER as quality characteristics of the digital signal from the output signals of both the equalizer and the Viterbi decoder. 
   The jitter controller  171  detects a jitter between the digital signal and the defined system clock. Furthermore, the jitter controller  171  outputs an enable signal when the cycle of the digital signal satisfies predetermined conditions, and then executes a predetermined calculation on the detected jitter according to the enable signal. These are fully described referring to  FIGS. 19 to 23 . 
     FIG. 19  is a view that illustrates a jitter controller according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 19 , the jitter controller according to an embodiment includes a jitter detector  1910 , a cycle examiner  1920 , and a jitter calculator  1930 . 
   The jitter detector  1910  and the cycle examiner  1920  receive the digital signals, respectively. The jitter detector  1910  detects a difference in time between the system clock outputted from the phase locked loop ( 130  shown in  FIG. 1 ) and the digital signal. When the digital signal is synchronous with the system clock, a zero crossing point of the digital signal with offset removed coincides with the system clock. However, in actual practice, a difference between the zero crossing point and the system clock signal occurs. The jitter detector  1910  computes the above difference in time. 
   Since a jitter can be detected only when a sign of the input signal shifts, an additional control signal representing a sign shift may be also outputted. Namely, sometimes there is a need for a control signal that shows whether the jitter is in a plus direction or a minus direction. 
   The cycle examiner  1920  finds a cycle by measuring a time length from a former sign&#39;s shift to a new sign&#39;s shift when the sign of the input signal shifts. Also the cycle examiner  1920  outputs an enable signal when the cycle satisfies a defined condition. A general example of the defined condition is ‘a case where the cycle is more than a fixed cycle’. In the above case, the enable signal is outputted only when the cycle of the input signals exceeds the fixed cycle so that a jitter calculation is executed only in that case. 
   Another example of the defined condition is to output the enable signal only when the cycle of the input signals is equal to a predetermined value. Additionally, in alternative examples, the enable signal may be outputted only when the cycle of the input signals is not equal to, more than, or not more than a predetermined value, or between predetermined values. 
   The jitter calculator  1930  executes a jitter calculation only when the cycle examiner  1920  satisfies a predetermined condition by using a jitter value outputted from the jitter detector  1910  and the output signal of the cycle examiner  1920 . That is, the jitter calculation is not always executed whenever the jitter happens. 
     FIG. 20  is a view that illustrates a jitter detector of the jitter controller according to an exemplary embodiment of the present invention. 
   Referring to  FIGS. 19 and 20 , the jitter detector  1910  may be embodied by analog techniques or digital techniques, but this specification discloses an example of digital techniques. 
   In the jitter detector  1910  using digital techniques, the input signal from which the offset is removed is inputted in the form of quantized digital data. Since the jitter is produced when the sign of the input signal shifts, the jitter detector  1910  may determine in advance whether the sign of the input signal shifts or not. 
   Specifically, one of the digital signals is delayed by one system clock through a delay unit  2010  and the other is not delayed. A first most significant bit extractor  2020  and a second most significant bit extractor  2030  detect most significant bits (MSBs) respectively from the delayed signal and the non-delayed signal. Since a sign shift causes a variation of the MSB, an XOR gate  2040  receiving the detected MSBs outputs “1” when the sign shift happens. Namely, an output without the sign shift is “0” and an output with the sign shift is “1”. So this signal may become a sign detection signal. 
   A method of computing the jitter is as follows. A first absolute value extractor  2050  and a second absolute value extractor  2060  receive the delayed signal and the non-delayed signal, respectively, and output their absolute values to a minimum value extractor  2070 . The minimum value extractor  2070  selects and outputs a relatively smaller absolute value. A divider  2080  divides the selected output signal by an added value of the absolute values of two input signals. Then an output value from the divider  2080  is inputted into an MUX  2090 . The MUX  2090  that normally outputs “0” selects the output signal of the divider  2080  when the sign shift happens. The MUX  2090  may multiply the output signal of the divider  2080  by a predetermined value. 
     FIG. 21  is a view that illustrates a calculation method of a jitter value according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 21 , referential characters “a” and “b” are values obtained by sampling the analog signal. The sum of referential characters “a′” and “b′” is fixed as the system clock. The jitter means a difference on a time axis between the clock signal and the input signal, and may be represented by a time difference between the zero crossing point of the input signal and the system clock. 
   Since a value of “b′” means a time difference between the system clock and the input signal, it may be jitter value. Supposing that a signal is linear around the zero crossing point, a relationship expression “a/a′=b/b′=(a+b)/(a′+b′)” is obtained. Here, since a, b, and a′+b′ (i.e., system clock) are already known values, the above expression can be rewritten as b′=b×(system clock)/(a+b). This calculation may be executed through the jitter detector shown in  FIG. 20 . As given above, the jitter value may be calculated by putting a smaller absolute data between two zero crossing points into a numerator and further putting an added value of two absolute values into a denominator. 
     FIG. 22  is a view that illustrates a cycle examiner of the jitter controller according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 22 , the cycle examiner of an exemplary embodiment includes an inverter  2210 , an edge detector  2220 , a reset counter  2230 , and a comparator  2240 . 
   The edge detector  2220  judges whether the input signal passes the zero point when the input signal is entered. As discussed above, the edge detector  2220  may detect the MSBs from both the delayed signal and the non-delayed signal, and outputs after an XOR operation of the detected MSBs. A value “1” is outputted at an edge point where the sign shift happens. When an edge is detected, the edge detector  2220  sends the counter  2230  a signal showing that the edge is detected. 
   The counter  2230  adds a count value whenever the system clock is entered. This clock may be an inversed clock passing through the inverter  2210 . 
     FIG. 23  is a view that illustrates a timing diagram of a counter of the cycle examiner according to an exemplary embodiment of the present invention. 
   Referring to  FIGS. 22 and 23 , when an edge detection signal is entered, the counter  2230  is reset to “0”. And a value just before reset is inputted into the comparator  2240 . The comparator  2240  outputs an enable signal by comparing an output value of the counter  2230  with a fixed value. A micro controller or other external controller may change in advance the fixed value. 
   The enable signal, i.e., the output signal of the comparator  2240 , may be used as an approval signal that decides whether to use the output of the cycle examiner or not. The comparator  2240  may be configured in a variety of forms. For example, the comparator  2240  may be configured to output the enable signal when a signal having a smaller cycle than a specific cycle is detected, when a signal having a greater cycle than a specific cycle is detected, when a signal having a specific cycle is detected, when a signal having a specific cycle is not detected, or when the cycle of a detected signal is between specific values. A micro controller or other external controller may regulate these cases. 
   Specifically, in case of input signals with 3T to 11T, the enable signal may be outputted when cycles with 3T to 11T are detected. In case of input signals with 3T to 11T and 14T, the enable signal may be outputted when cycles with 3T to 11T and 14T are detected. In case of input signals with 2T to 9T, the enable signal may be outputted when cycles with 2T to 9T are detected. 
   Furthermore, in case of input signals with 3T to 11T, the enable signal may be outputted when cycles with 4T to 11T are detected. In case of input signals with 3T to 11T and 14T, the enable signal may be outputted when cycles with 4T to 11T and 14T are detected. In case of input signals with 2T to 9T, the enable signal may be outputted when cycles with 3T to 9T are detected. 
   Additionally, in case of input signals with 3T to 11T, the enable signal may be outputted when cycles more than 3T are detected. In case of input signals with 3T to 11T and 14T, the enable signal may be outputted when cycles more than 3T are detected. In case of input signals with 2T to 9T, the enable signal may be outputted when cycles more than 2T are detected. 
   Additionally, in case of input signals with 3T to 11T, the enable signal may be outputted when cycles more than 4T are detected. In case of input signals with 3T to 11T and 14T, the enable signal may be outputted when cycles more than 4T are detected. In case of input signals with 2T to 9T, the enable signal may be outputted when cycles more than 3T are detected. 
   Returning to  FIG. 19 , the jitter calculator  1930  receives the detected jitter value and the approval signal of the cycle examiner  1920  and then executes a calculation of a real jitter value. The jitter calculator  1930  outputs an average of the jitters in specific cycles when predetermined conditions are satisfied. The cycle examiner  1920  may establish these conditions. Generally the real jitter value is an average of jitter values obtained within specific cycles, and computed by a micro controller or any other arithmetic hardware under particular conditions that a system designer requires. ‘To output an average jitter value when a signal more than 4T happens n times’ is an example of such conditions. Generally possible conditions are as follows. 
   To output an average jitter value when a specific T happens N times. 
   To output an average jitter value when a cycle more than a specific T happens N times. 
   To output an average jitter value when a cycle less than a specific T happens N times. 
   To output an average jitter value when a cycle between T 1  and T 2  happens N times. 
   To output an average jitter value when a non-T signal happens N times. 
   To output an average jitter value for every specific time. 
   Here, T, T 1 , T 2 , N are optionally selectable values by a micro controller or any other controller. Furthermore, the above conditions may be variously established by changing the conditions of the cycle examiner  1920 . 
   The aforesaid jitter controller  171  may be embodied to involve the configuration and the operation of “apparatus and method of jitter detection” disclosed in the Korean Patent Publication No. 2004-0099951. 
   Returning again to  FIG. 1 , the SbER controller  172  of the signal quality measurer  170  computes quality characteristics (i.e., SbER) of the digital signal by adding products of a probability (C T ) that a pattern T of the digital signal happens, a probability (erf(0)) that the pattern T is detected corresponding to a pattern F of the digital signal, and a Hamming distance between the pattern T and the pattern F. This is discussed hereinafter referring to  FIGS. 17 and 18 . 
     FIG. 17  is a block diagram that illustrates an SbER controller of a signal quality measurer according to an exemplary embodiment of the present invention. 
   Referring to  FIG. 17 , the SbER controller  1700  of an exemplary embodiment includes a register  1710 , a pattern comparator  1720 , a reference table  1730 , and a calculator  1740 . 
   The equalized input signal is inputted into the Viterbi decoder. Furthermore, the input signal is transmitted to the SbER controller  1700  and recorded into the register  1710 . 
   Binary data outputted from the Viterbi decoder are inputted into the pattern comparator  1720 . The pattern comparator  1720  compares patterns of the binary data with the reference table  1730  exemplarily shown in  FIG. 18 . Thereafter, the calculator  1740  calculates a difference between two Euclidean distances given below, computes the Hamming distance (H T, F ), and computes the SbER using the following equation 3. 
   Equalized input signal—pattern T in  FIG. 18  (right pattern) 
   Equalized input signal—pattern F in  FIG. 18  (erroneous pattern)
 
 SbER=ΣC   T   *erf (0)* H   T, F   [Equation 3]
         C T : A probability that a pattern T of the digital signal happens.   erf(0): A probability that the pattern T is detected corresponding to a pattern F of the digital signal.   H T, F : A Hamming distance between the pattern T and the pattern F.       

   By computing the SbER through the Equation 3, the SbER controller  1700  can measure quality characteristics of the digital signal. The aforesaid SbER controller  1700  may be embodied to involve the configuration and the operation of “apparatus and method of jitter detection” disclosed in the Korean Patent Publication No. 2004-0099951 and those of “signal quality evaluation method, information recording and reproducing system, recording compensation method, and information medium” disclosed in the Japanese Patent Publication No. 2003-151219. 
   As fully discussed hereinbefore, the optical disc reproducing apparatus of the present invention is configured with a combination of the asymmetry compensator, the phase locked loop, the binary module, the equalizer, the adaptive level error detector, and the signal quality measurer. A disc reproduction realized by combining featured configurations and operations of the above elements may create a synergy effect. 
   Specifically, the optical disc reproducing apparatus of an exemplary embodiment of the present invention may allow a disc reproduction supporting both RLL(1,10) and RLL(2,10) codes. 
   Furthermore, the optical disc reproducing apparatus of an exemplary embodiment of the present invention may allow an effective disc reproduction under a high-density environment more than 15 GB in a 12 cm diameter disc by supporting a 5-Tap PRML structure. 
   Additionally, the optical disc reproducing apparatus of an exemplary embodiment of the present invention may effectively detect frequency errors under a high-density environment with a frequent ISI and a high noise. 
   Additionally, the optical disc reproducing apparatus of an exemplary embodiment of the present invention may execute an asymmetric compensating operation in response to an RLL(1,10) environment. 
   Additionally, the optical disc reproducing apparatus of an exemplary embodiment of the present invention may execute a minimum T signal compensating operation under 3T and 2T environments and in response to an RLL(1,0) code. 
   Although a few exemplary embodiments of the present invention have been shown and described, the present invention is not limited to the described exemplary embodiments. Instead, it would be appreciated by those skilled in the art that changes may be made to these exemplary embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the claims and their equivalents.