Patent Publication Number: US-11399236-B2

Title: Sensor device and microphone assembly

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This patent application claims the benefit of and priority to U.S. Provisional Application No. 62/613,738, filed Jan. 4, 2018 and PCT Application Number PCT/US2018/068206 filed Dec. 31, 2019, the content of which are incorporated herein by reference in their entirety. 
    
    
     FIELD OF THE DISCLOSURE 
     The present disclosure relates generally to microphones, and more particularly to sensing low frequency atmospheric pressure. 
     BACKGROUND 
     Several audio sensing applications include electronic microphones. Some such microphones include microelectromechanical systems (MEMS) microphones, e.g., capacitive microphones, the capacitance of which changes as a function of incident changes in pressure. The microphones transform the change in capacitance into corresponding electrical signals. It is desirable to sense a wide range of frequencies. In some implementations, it may be desired to sense both acoustic signals as well as low-frequency pressure signals such as atmospheric or ambient pressure. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  shows a block diagram of a first example microphone device in accordance with various embodiments. 
         FIG. 1B  shows an example electrical signal generated by the first example microphone device shown in  FIG. 1A  in accordance with various embodiments. 
         FIG. 2  shows additional details of the digital-to-analog converter shown in  FIG. 1A  in accordance with various embodiments. 
         FIG. 3  shows a block diagram of an exemplary digital loop filter in accordance with various embodiments. 
         FIG. 4  shows a block diagram of an example Pulse-Width and Pulse-Amplitude Modulator (PWAM) in accordance with various embodiments. 
         FIG. 5  shows a block diagram of an example noise-shaping up-sampler and quantizer in accordance with various embodiments. 
         FIG. 6  shows an example block diagram of the operation of a modulator portion of a hybrid Pulse-Width and Pulse-Amplitude Modulator (PWAM) in accordance with various embodiments. 
         FIG. 7  shows a schematic block diagram of an example current output converter in accordance with various embodiments. 
         FIG. 8  shows a schematic block diagram of an example controllable current generator IDACN of the current output converter in accordance with various embodiments. 
         FIG. 9A  shows a schematic block diagram of the controllable current generator IDACN of the current output converter when placed in a first state in accordance with various embodiments. 
         FIG. 9B  shows a schematic block diagram of the controllable current generator IDACN of the current output converter  130  when placed in a second state in accordance with various embodiments. 
         FIG. 10  shows an example graph of the sensitivity of a capacitive transducer at two different temperatures in accordance with various embodiments. 
         FIG. 11  shows a second example microphone device having a command and control circuit in accordance with various embodiments. 
         FIG. 12  shows an exemplary embodiment of a microphone assembly or system in accordance with various embodiments. 
         FIG. 13  shows a block diagram of a third example microphone device in accordance with various embodiments. 
         FIG. 14  shows a block diagram of a cross-over filter circuit shown in  FIG. 13  in accordance with various embodiments. 
         FIG. 15  shows an exemplary embodiment of a microphone assembly or system in accordance with various embodiments. 
         FIG. 16  shows a flow diagram of a process for generating a low frequency pressure signal and an audio signal in a microphone device in accordance with various embodiments. 
     
    
    
     DETAILED DESCRIPTION 
     The present disclosure describes devices and techniques for sensing low-frequency pressure as well as acoustic signals using the same microphone device. The microphone device can include a transducer that is configured to sense pressure changes in an audible or acoustic frequency range as well as at lower frequencies. The transducer can transform the sensed pressure into electrical signals. An electrical circuit can process the electrical signals generated by the transducer and provide an audio signal corresponding to the pressure changes in the audible frequency range and a low frequency pressure signal corresponding to low frequency pressure changes. In some embodiments, the transducer can be implemented using microelectromechanical systems (MEMS) technology or using electret material. In other embodiments, the transducer is embodied as a MEMS piezoelectric, a capacitive transducer, or other transducer. In some embodiments, the transducer can be configured to sense pressure changes in a frequency range of about 1 Hz to about 20 kHz. In some embodiments, the transducer can be configured to sense pressure changes between about 0.1 Pa to about 1 Pa. 
     In some embodiments, the microphone device, in addition to the transducer, can include a pressure transducer for measuring absolute pressure and pressure changes at frequencies below the frequency range of the transducer. For example, the pressure transducer can be utilized to measure absolute pressure and pressure changes at frequencies in the range from about 0 Hz to about 1 Hz. In some embodiments, the microphone device can combine the electrical signal from the pressure transducer and the low frequency pressure signal generated by the pressure transducer. 
     The electrical circuit can include a first processing circuit for generating the audio signal and a second processing circuit for generating the low frequency pressure signal. The first processing circuit can include a forward path circuit, a feedback circuit, and a summing node. At the summing node, a feedback signal from the feedback circuit can be used to remove low frequency components from the electrical signal generated by the transducer before the electrical signal is applied to the first processing circuit. The forward path circuit can amplify the electrical signal processed by the canceling circuit to generate the audio signal. The feedback circuit can include a low pass filter that filters the audio signal to generate a low pass filter signal. The feedback circuit can generate the feedback signal based on the low pass filter signal and provide the feedback signal to the summing node. The second processing circuit can process the low pass filter signal to generate the low frequency pressure signal. The first processing circuit and/or second processing circuit may be an integrated circuit implemented as an Application Specific Integrated Circuit (ASIC). 
     In some embodiments, the second processing circuit, in generating the low frequency pressure signal, can process the low pass filter signal to compensate for non-linearity of the transducer. In some embodiments, the second processing circuit is configured to generate the low frequency pressure signal based on both the low pass filter signal and the audio signal. In some embodiments, the transducer can include a high pass filter having a cut-off frequency below the audible frequency range. In some such embodiments, the second processing circuit can generate the low frequency pressure signal based on compensation for a reduced sensitivity of the transducer below the cut-off frequency. In some embodiments, the canceling circuit can include a summing circuit for subtracting the feedback signal from the electrical signal generated by the transducer before the electrical signal is provided to the forward path circuit. In some other embodiments, the canceling circuit can include a charge pump, coupled to the transducer, where the charge pump can generate charge flow based on the low pass filter signal such that low frequency components in the electrical signal generated by the transducer are attenuated or suppressed. 
     The forward path circuit can include an analog-to-digital converter (ADC) that can convert the analog audio signal into a digital audio signal. In some embodiments, the low pass filter can filter the digital audio signal to generate a low pass filter signal that also is digital. The feedback circuit can include a digital-to-analog converter (DAC) that can convert the digital low pass filter signal into an analog feedback signal, and provide the analog feedback signal to the canceling circuit. 
     In some embodiments, the microphone device can include a second pressure transducer used to measure absolute pressure. Generally, for a single pressure transducer to measure absolute pressure, the pressure transducer should have a stiff diaphragm. However, the stiff diaphragm makes small pressure change measurements difficult. To allow small pressure change measurements, the stiff diaphragm may be increased in size, thereby increasing the overall size of the pressure sensor and the package in which it is housed. In the microphone device discussed above, the small pressure changes are instead measured by the transducer, which also measures pressure changes in the audible frequency range. Thus, the second pressure transducer can be dedicated to measuring absolute pressure changes, and can therefore have a relatively smaller size. 
       FIG. 1A  shows a block diagram of a first example microphone device  100 . The microphone device  100  includes a transducer  101 , a first processing circuit  102 , and a second processing circuit  104 . The first processing circuit  102  can include a forward path circuit  106 , a feedback path circuit  108 , and a summing node  110 . The second processing circuit  104  includes a compensation circuit  112 . The transducer  101  can be configured to sense pressure changes in an audible or acoustic frequency range as well as at lower frequencies. The transducer  101  can transform the sensed pressure into electrical signals. In some embodiments, the transducer  101  can be implemented using microelectromechanical systems (MEMS) technology or using electret material. In other embodiments, other transducer technologies can be used; the present disclosure is not limited to application using MEMS sensors. In some embodiments, the transducer  101  can be configured to sense pressure changes in a frequency range of about 1 Hz to about 20 kHz, or about 0.1 Hz to about 20 kHz. The transducer can be configured to sense pressure changes that have an amplitude below 1 Pa. The first processing circuit  102  and/or second processing circuit  104  may be implemented within an ASIC, in some implementations. 
     The transducer  101  can generate an electrical signal  122  based on sensed pressure. In particular, the electrical signal  122  can include frequency components corresponding to a first frequency range and a second frequency range, where the first frequency range can correspond to a sub-acoustic frequency range, or a frequency range below a human-audible frequency range, and the second frequency range includes frequencies in the human audible frequency range. For example, the first frequency range can include frequency components having frequencies between about 1 Hz to about 20 Hz or about 0.1 Hz to about 20 Hz, and the second frequency range can include frequency components having frequencies between 20 Hz to about 20 kHz. In some embodiments, the transducer can be implemented using microelectromechanical systems (MEMS) technology or using electret material. In other embodiments, the transducer is embodied as a MEMS piezoelectric or other transducer. In some embodiments, the transducer  101  can be a capacitive transducer. 
       FIG. 1B  shows an example electrical signal  400 . For example, the example electrical signal  400  can represent the electrical signal  122  generated by the transducer  101  shown in  FIG. 1A . The example electrical signal  400  corresponds to pressure changes sensed by a transducer, such as the transducer  101 . The pressure changes can include low frequency pressure changes and high frequency pressure changes. The example electrical signal  400  can include a superposition of the frequency components corresponding to these pressure changes. For example, the example electrical signal  400  includes a first frequency component having a first frequency  1 /λ 1  and a second frequency component  1 /λ 2 . The first frequency  1 /λ 1 . The first frequency  1 /λ 1  is less than the second frequency  1 /λ 2 . For example, the first frequency  1 /λ 1  can be within the first frequency range discussed above, while the second frequency  1 /λ 2  can be within the second frequency range discussed above. 
     Referring again to  FIG. 1A , the electrical signal  122  is provided to the summing node  110 , which precedes the forward path circuit  106 . In some embodiments, low frequency components in the electrical signal may saturate amplification circuitry in the forward path circuit  106 . To reduce the risk of saturation, the low frequency components in the electrical signal  122  can be attenuated before the electrical signal is fed to the forward path circuit  106 . The summing node  110  can receive a feedback signal  124 , which includes low frequency components within the first frequency range. The low frequency components can correspond to the low frequency components generated by the transducer  101  (e.g., corresponding to ambient pressure). At the summing node  110  the low frequency components can be removed from the electrical signal  122 , thereby resulting in attenuation of the low frequency components in the audio signal, which is provided to an input of the forward path circuit  106 . In one or more embodiments, the low frequency components can be attenuated by about 30 dB to about 60 dB. 
     The forward path circuit  106  can include an amplifier  114  and an analog-to-digital converter (ADC)  116 . The amplifier  114  can amplify the amplitude of the audio signal and the ADC  116  can convert the amplified audio signal into a digital audio signal, and provide the digital audio signal at an output of the forward path circuit  106 . In some embodiments, the forward path circuit  106  may not include the ADC  116 , and can instead provide an analog audio signal at its output. In some embodiments, the forward path circuit  106  also can include filters, level shifters, and/or other processing circuitry for conditioning the audio signal. 
     The feedback path circuit  108  can include a loop filter (LF)  118  and a digital-to-analog converter (DAC). The LF  118  can include, for example, a low pass filter having a cut-off frequency that can filter the audio signal  127  such that it passes frequency components below the cut-off frequency while blocking frequency component above the cut-off frequency. That is, the LF  118  can generate a low pass filter signal that includes low pass frequency components within the audio signal. In some embodiments, the cut-off frequency can correspond to the lowest frequency in the second frequency range. For example, the cut-off frequency can be about 20 Hz. In some embodiments, the LF  118  can be programmable/customizable, such that the cut-off frequency and the amplitude and phase response of the LF  118  can be changed. In instances where a digital low pass filter is used to implement the LF  118 , the values of the coefficients of the digital low pass filter can be changed to change the characteristics, such as the cut-off frequency, the amplitude response, and the phase response, of the LF  118 . In one or more embodiments, in particular where the transducer  101  is a capacitive transducer, the LF  118  can have a transfer function or a difference equation that is a constant. Such an LF  118  in combination with the capacitive transducer can form a low pass filter. 
     The low pass filter signal  126  includes low frequency components of substantially the same frequency and amplitude as the low frequency components generated by the transducer  101 . The DAC  120  can convert the digital low pass filter signal  126  into an analog feedback signal  124 , and provide the analog feedback signal  124  to the summing node  110 . 
       FIG. 2  shows additional details of the DAC  120  shown in  FIG. 1A . In particular, the DAC  120  can include a hybrid Pulse-Width and Pulse-Amplitude Modulator (PWAM)  125  connected in series with a current output converter (IDAC)  130 . The DAC  120  is configured to convert the low pass filter signal  126  into the corresponding analog feedback signal  124 , which is applied to summing node  110 . A transducer  101  output may have a high impedance, e.g. a capacitance of about 0.5 pF to about 10 pF. This property of transducers, and the design and electrical properties, in particular output impedance, of the current output converter  130  are discussed in further detail below with reference to the schematic diagram of the converter  130 . The application of the analog feedback signal  124  to the transducer  101  output leads to numerous advantages compared with prior art approaches. The direct coupling of the analog feedback signal  124  to the transducer  101  output effectively prevents low-frequency overload of the forward path circuit  106 . This is accomplished by the low-pass filtering of the analog feedback signal  124  carried out by the digital loop filter  118 , which low-pass filtering cancels or suppresses low-frequency components of the electrical signal  122  at the summing node  110 , or input nodes, of the amplifier  114 . Furthermore, the noise floor of the microphone device  100  may be lowered by tailoring a frequency response of the transducer  101  to the accurate frequency response of the forward path circuit  106 . Furthermore, the accurate control over the frequency response of the forward path circuit  106  improves frequency response matching, including phase matching, between individual microphone devices of a beamforming microphone array which may include two, three, or more microphone devices. This improved response matching leads to improved, predictable and stable directional response of the beamforming microphone array. 
     The upper portion of  FIG. 3  shows a block diagram of an exemplary embodiment of the previously discussed digital loop filter  118  of the feedback path circuit  108 . The digital loop filter  118  has second order low pass filter characteristics using a classical IIR filter bi-quad topology. The skilled person will understand that other digital filter types and topologies, such as FIR filters or other types of IIR filter topologies, may be utilized in alternative embodiments of the digital loop filter  118 . Likewise, other filter orders may be used. The transfer function of the illustrated digital loop filter  118  is determined by values of the filter coefficients which include: a1, a2, b0, b1 and b2. The frequency response graph  300  of the lower portion of  FIG. 3  shows an exemplary magnitude response  310  of the digital loop filter  118  where the low pass cut-off frequency has been tuned to about 200 Hz. The corresponding magnitude response  305  of the forward path circuit  106  for this particular setting of the response  310  of the digital loop filter  118  is also plotted. The skilled person will notice the expected 2 nd  order highpass magnitude response of the forward path circuit  106 , with a highpass cut-off frequency set to approximately 30 Hz. The skilled person will understand that the lowpass cut-off frequency of the digital loop filter  118  may be adjusted over a broad frequency range to obtain a desired highpass cut-off frequency of the forward path circuit  106 . The latter highpass cut-off frequency may be situated in the frequency range between about 1 Hz and about 200 Hz for various embodiments of the microphone device depending on requirements of a specific application. In one or more embodiments, the digital loop filter  118  can be a low pass filter of various orders. For example, the digital loop filter  118  can be a first order low pass filter with one pole and one zero. 
     The skilled person will understand that certain embodiments of the first processing circuit  102  may include an adjustable or programmable transfer function of the digital loop filter  118  where the transfer function is controlled by filter configuration data. The filter configuration data may include respective values of one or more of the previously discussed filter coefficients a1, a2, b0, b1 and b2. The filter configuration data may be received by the first processing circuit  102  via a command and control interface from a host processor. The programmable transfer function of the digital loop filter  118  allows the microphone assembly to be tailored to requirements of a particular application after manufacturing in a flexible manner and therefore reduces the number of variants needed of the microphone assembly. 
     Other types of configuration data for various circuits and functions of the microphone device  100  may likewise be programmed through the command and control interface. The configuration data, including filter configuration data, may be stored in rewriteable memory cells (not shown) of the processing circuit, such as flash memory, EEPROM, RAM, register files or flip-flops. These rewriteable memory cells may hold or store certain default values of the filter configuration data. 
       FIG. 4  shows a block diagram of an example Pulse-Width and Pulse-Amplitude Modulator (PWAM)  125 . The output of the digital loop filter  118  is connected to the input of the PWAM  125  such that the previously discussed low pass filter signal  126  is applied to the input of the PWAM  125 . The low pass filter signal  126  may be a multibit signal with a relatively high resolution—for example between 16 and 32 bits per sample, such as 24 bits per sample, to maintain a high signal resolution through the feedback path circuit  108 . The sampling frequency of the first digital feedback signal may lie between 32 kHz and 384 kHz, for example between 96 kHz and 192 kHz. The PWAM  125  includes a noise-shaping up-sampler and quantizer  410  at the input receiving the low pass filter signal  126 . The noise-shaping up-sampler and quantizer  410  raises the sampling frequency of the low pass filter signal  126  with a pre-set or programmable ratio—for example an integer ratio between 2 and 16 to generate a digital feedback signal  415  at a second sampling frequency. The noise-shaping up-sampler and quantizer  410  is furthermore configured to quantize samples of the digital feedback signal  415  to a smaller number of bits than the samples of the low pass filter signal  126 . According to one exemplary embodiment of the quantizer  410  the samples of the low pass filter signal  126  has 24 bits per sample while the samples of the digital feedback signal have  415  been decimated to 11 bits. These samples may be generated according to a signed sample format where a sign bit takes one bit and a magnitude portion is represented by the residual 10 bits of the sample. 
       FIG. 5  shows a block diagram of an example noise-shaping up-sampler and quantizer  410  of the PWAM  125 . The low pass filter signal  126  is represented by X(z) and the digital feedback signal by X(z)+E(z), where E(z) represents a quantization noise component caused by the quantization operation carried out by the quantizer  504 . The noise-shaping up-sampler and quantizer  410  includes a noise-shaping feedback loop extending through loop filter Hn(z)  506  to a second adder  510  on the input side, which adder shapes the spectrum of the generated quantization noise to higher frequencies and therefore maintains a relatively high resolution of the digital feedback signal throughout the audio frequency range despite the quantization. The noise-shaping up-sampler and quantizer  410  may include a feedforward loop as illustrated extending to an output side summer  512 . The noise-shaping up-sampler and quantizer  410  may also include the illustrated Dither generator  502 , which adds a pseudo-random noise signal of appropriate level to the first digital feedback signal at the input of the noise-shaping up-sampler and quantizer  410  using a first input side adder  508 . This pseudo-random noise signal may reduce audible artefacts associated with the quantization operation in a well-known manner. 
     Referring again to  FIG. 4 , the PWAM  125  additionally includes a modulator  420  connected to the output of the noise-shaping up-sampler and quantizer  410  for receipt of the digital feedback signal  415  (X(z)+E(z)). The operation and functionality of an exemplary embodiment of the modulator  420  is schematically illustrated in  FIG. 6 . The modulator  420  takes the digital feedback signal (X(z)+E(z)) in the multibit (PCM) format and converts the digital feedback signal into a pulse-width and pulse-amplitude modulated signal. The sampling frequency of this pulse-width and pulse-amplitude modulated signal may be markedly higher than the sampling frequency of the digital feedback signal as discussed below. The sampling frequency of the pulse-width and pulse-amplitude modulated signal  425  may be at least 16 times higher than the sampling frequency of the digital feedback signal, such as 32 or 64 times higher. One embodiment of the modulator  420  accepts a 192 kHz sampling frequency of the digital feedback signal and generates a corresponding pulse-width and pulse-amplitude modulated signal at a sampling frequency of 12.288 MHz, and hence raises the sampling frequency of the latter by an upsampling factor of 64. The pulse-width and pulse-amplitude modulated signal  425  may be applied to a current converter control circuit or block  430 —(see, e.g.,  FIG. 4 ). The current converter control circuit  430  is configured to convert or transform the pulse-width and pulse-amplitude modulated signal into a corresponding sequence of variable width and amplitude current pulses at the output of the previously discussed current output converter (IDAC)  130  by controlling how individually controllable current generators (illustrated in  FIGS. 7 and 8 ) are activated. The current converter control circuit  430  may include an appropriately configured digital state machine. One embodiment of the current converter control circuit  430  may include a dynamic element matching circuit  432  as schematically illustrated where the selection of individually controllable current generators of the current output converter is carried out in a randomized manner to average out offsets between nominally identical current generators. 
     Referring again to  FIG. 6 , the digital feedback signal X(z)+E(z) is applied to the input of the modulator  420  and the sampling frequency raised with a predetermined ratio, N, such as 64. In the present embodiment, the resolution of the second digital feedback signal is 11 bits as discussed previously. A dividing block or circuit  603  divides each 11 bits sample of the digital feedback signal with N to compute respective modulus values and remainder values of the samples. The drawing shows four exemplary values,  138 ,  40 ,  522  and  276  using decimal notation, of the 11 bit samples of the digital feedback signal expressed in decimal format initially. The decimal sample value  138  is divided by 64 producing a modulus value of 2 and a remainder value of 10 as illustrated. The corresponding computation is also illustrated for the three remaining samples  40 ,  522  and  276 . The decimal sample value  138  is converted into binary format showing how the modulus value 2 corresponds to 00010b and the remainder value of 10 corresponds to 001010b. A first variable width and amplitude pulse  610  of the pulse-width and pulse-amplitude modulated signal is generated by conversion of the decimal sample value  138 . The first variable width and amplitude pulse  610  is essentially constructed from two segments. A first pulse segment (2*64) has an amplitude of “2” (y-axis scale) spanning over a full pulse width, i.e. 100% modulation and pulse amplitude of 2—hence representing the modulus value “2.” The first variable width and amplitude pulse  610  additionally includes a second pulse segment (1*10) spanning over merely 10 sample time clocks of the 12.288 MHz sampling frequency of the pulse-width and pulse-amplitude modulated signal. Stated in another way, the decimal sample value  138  is converted into an “analog” variable width and amplitude pulse with a corresponding pulse area. 
     The conversion of the decimal sample value  40  into the second variable width and amplitude pulse  620  is also illustrated. The decimal sample value  40  leads to a modulus value of 0 and a remainder value of 40 as illustrated. The corresponding, second, variable width and amplitude pulse  620  reflects this outcome by merely including a second pulse segment (1*40) with a “one” amplitude and spanning over merely 40 sample time clocks of the 12.288 MHz sampling frequency of the pulse-width and pulse-amplitude modulated signal. The conversion of the decimal sample value  522  into a third variable width and amplitude pulse  630  is finally illustrated using the same principles outlined above. The skilled person will understand that the modulator  420  is configured to convert incoming sample values into corresponding sequences of variable width and amplitude pulses where the pulse area of each of the variable width and amplitude pulses  610 ,  620 ,  630  represents the sample value in question. Hence, each of the variable width and amplitude pulses  610 ,  620 ,  630  can be viewed as an analog representation of the sample value in question. 
     The skilled person will understand that the modulator  420  may be configured to generate the variable width and amplitude pulses following different modulation schemes. In the present embodiment, each of the variable width and amplitude pulses is preferably centered at a midpoint of the pulse period, e.g., centered at the sample clock time  32  in this embodiment using an upsampling factor of 64. This centering is often referred to as a double-edge pulse-width modulation. However, other embodiments of the modulator  420  may be adapted to build the variable width and amplitude pulses by applying single-edge modulation. 
       FIG. 7  shows a schematic block diagram of an example current output converter  130 . The current output converter  130  includes a predetermined number, N, of individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , . . . IDACN, for example between 4 and 32 current generators, such as 16 current generators. The respective outputs of the N individually controllable current generators are connected in parallel to a common DAC output node  131 . A capacitive transducer  702  is connected to the common DAC output node  131 , which, in turn, can be connected to the summing node  110 . The skilled person will understand that the capacitive transducer  702  may include the previously discussed transducer  101  of the microphone device  100  shown in  FIG. 1A . However, other types of capacitive transducer elements for sensing of various types of physical variables may in the alternative be driven by the present current mode DAC. The N individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN may be nominally identical, but the skilled person will understand that component variations associated with semiconductor manufacturing may cause minor variations of characteristics between the controllable current generators, in particular current sinking and sourcing capabilities. Each of the N individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN is configured to selectively source current into the capacitive transducer  702  or sink current from the capacitive transducer  702  in accordance with the switching control carried out by the current converter control circuit  430 , and thereby charge or discharge voltage across the capacitive transducer  702 . Each of the N individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN can be considered a one-bit or 1.5 bit binary values +1 or −1. The sourcing and sinking of the predetermined current amount or level may be carried out by selecting between a first state and second state of the controllable current generator. Finally, each of the individually controllable current generators may include a third state or an idle/zero output state where the current generator neither sources nor sinks current to/from its output. In this idle state, the current generator may be placed in a high-impedance mode effectively disconnecting the current generator from the common DAC output  131  as discussed in further detail below. The skilled person will appreciate that the maximum positive output value of the current converter may correspond to setting all N individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN to source current while the maximum negative output value corresponds to setting all N individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN to sink current. 
       FIG. 8  shows a schematic block diagram of an example controllable current generator IDACN of the current output converter  130  when placed in the idle state or off-state discussed above. The controllable current generator IDACN includes a first current source  802  and a second current source  804  connected in series between the positive DC supply rail VDD and a negative DC supply rail which is ground (GND) in the present embodiment. A first switch pair, including switches SW 2  and SW 5 , is coupled in-between the first and second current sources  802 ,  804  and is operating in a synchronized manner where both switches are simultaneously closed/conducting or open/non-conducting. The switches of the first switch pair SW 2  and SW 5  are closed in the idle state while the residual SW 1 , SW 3 , SW 4  and SW 6  are placed in open/non-conducting states as illustrated. This means that the current flowing through the first current source  802  and second current source  804  runs directly from VDD to GND as illustrated by the current path  810 . Consequently, each of the first and second current sources  802 ,  804  is electrically disconnected from the output node  831  and the controllable current generator IDACN does therefore not source or sink any noticeable current to the capacitive transducer  702  when placed in the idle state. 
     The controllable current generator IDACN additionally includes a DC voltage reference  806  connected to an inverting input of a differential loop amplifier  808 , e.g. an operational amplifier or other differential amplifier, of a feedback regulation loop of the IDACN. The voltage of the DC voltage reference  806  may be equal to one-half VDD. The differential loop amplifier  808  has a non-inverting input (+) connected to a midpoint node  812  arranged in-between the first switch pair SW 2 , SW 5 . An output of the differential loop amplifier  808  is connected to a control input  805  of the second current source  804  where the control input  805  is configured to adjust the current level of the second current source  804 . The operation of the differential loop amplifier  808  therefore seeks to dynamically or adaptively adjust the voltage at the midpoint node  812  to approximately one-half VDD, which is the voltage set at the negative input of the differential loop amplifier  808 , by adjusting the current flowing through second current source  804  via the control input  805 . This adaptive adjustment of the voltage at the midpoint node  812  is carried out by a feedback regulation loop. Hence, the differential loop amplifier  808 , the second current source  804  and the DC voltage reference  806  therefore jointly form a DC error suppression circuit which is configured to match or align the first and second current levels supplied by the first and second current sources  802 ,  804  during the idle state of the controllable current generator IDACN. In certain embodiments, the differential loop amplifier  808  may possess a relatively small bandwidth, or large time constant, compared to the sampling frequency of the incoming pulse-width and pulse-amplitude modulated signal. The upper cut-off frequency of the differential loop amplifier  808  may, for example, be smaller than 100 kHz, or smaller than 40 kHz, which effectively performs a slow averaging of the current source balancing to secure a long-term zero DC offset at the output of each of the controllable current generators. 
     This property has several noticeable advantages, for example leading to a linear I/O characteristic of the current output converter  130 . The DC error suppression circuit also prevents build-up of DC voltage components on the load, which is a noticeable advantage in connection with driving transducer elements (such as, for example, capacitive transducers) where DC off-sets or DC imbalances of the analog feedback signal will tend to drive the DC operating point of the transducer away from a target DC operating point. This potential build-up of DC off-set is caused by the charge integration carried out by the capacitance of the transducer element. The controllable current generator IDACN is operating in the previously discussed idle state where the output node  831  is in a high-impedance state supplying substantially zero current output. Each of the switches SW 1 , SW 2 , SW 3 , SW 4 , SW 5  and SW 6  may include a controllable semiconductor switch for example a MOSFET. Each of the switches SW 1 , SW 2 , SW 3 , SW 4 , SW 5  and SW 6  may include a control terminal, for example a gate terminal of a MOSFET, which switches the controllable semiconductor switch between its conducting and non-conducting states. These control terminals are connected to the previously discussed current converter control circuit  430 . The current level supplied by the first and second current sources  802 ,  804  may vary depending on requirements of a particular application such as a load impedance, e.g. the capacitance of the capacitive transducer  702  in the present embodiment, the sampling frequency of the pulse-width and pulse-amplitude modulated signal, the number of parallel connected controllable current generators of the current output converter  130 , etc. In one exemplary embodiment of the current output converter  130  comprising 16 controllable current generators, the respective currents of the first and second current sources  802 ,  804  are set to about 100 pA, e.g. between 50 pA and 200 pA, when configured for driving a 1-4 pF capacitive transducer. The current settings of the controllable current generators generally depend on a dv/dt at the peak amplitude of the analog feedback signal at the highest frequency of interest of the feedback loop. The currents of the controllable current generators should preferably be capable of charging the capacitance of the transducer  101  without slew-induced distortion under these conditions. The highest frequency of interest of the analog feedback signal may lie between 300 Hz and 3 kHz, for example about 1 kHz, in exemplary embodiments of the microphone device  100 . An output impedance at 10 kHz of each of the individually controllable current generators IDAC 1 , IDAC 2 , IDAC 3 , IDACN is in some embodiments preferably larger than 1 MΩ, such as larger than 10 MΩ or 100 MΩ, when operating in either the first state or the second state. 
       FIG. 9A  shows a schematic block diagram of the controllable current generator IDACN of the current output converter  130  when placed in the first state, or +1 state, discussed above where the output  831  is sourcing the predetermined current level to the capacitive transducer  702  or other load circuit. In the first state, the switches of the first switch pair SW 2  and SW 5  are open or non-conducting and the switches SW 1  and SW 6  are both open or non-conducting as illustrated. The residual switches SW 4  and SW 3  are in contrast placed in conducting or closed states as illustrated. This combination of switch states means that the current flowing through the first current source  802  is sourced into the capacitive transducer  702  via current path  810   a  while the current generated by the second current source  804  runs from the DC voltage reference  806 , which may be equal to one-half VDD, directly to GND via the current path  810   b . Consequently, the controllable current generator IDACN sources the predetermined current level to the capacitive transducer  702  when placed in the first state. The skilled person will understand that the DC balancing of the current levels of the first current source  802  and the second current source  804  is still maintained by the operation of the previously discussed DC error suppression circuit. 
       FIG. 9B  shows a schematic block diagram of the controllable current generator IDACN of the current output converter  130  when placed in the second state, or −1 state, discussed above where the output  831  is sinking the predetermined current level from the capacitive transducer  702  or other load circuit to discharge the load circuit. In the second state, the switches of the first switch pair SW 2  and SW 5  are open or non-conducting and the switches SW 4  and SW 3  are both open or non-conducting as illustrated. The residual switches SW 1  and SW 6  are in contrast placed in conducting or closed states as illustrated. This combination of switch states means that the current flowing through the first current source  802  is sourced into the DC voltage reference  806  and thereafter to GND via the current path  880   a . In contrast, the predetermined current generated by the second current source  804  is drawn out of the capacitive transducer  702  via current path  880   b  to discharge the capacitive transducer  702 . Consequently, the controllable current generator IDACN sinks the predetermined current level from the capacitive transducer  702  when placed in the second state. The skilled person will understand that the DC balancing of the current levels of the first current source  802  and the second current source  804  is still maintained by the operation of the previously discussed DC error suppression circuit. 
     The skilled person will understand that above outlined switch arrangement and associated switching scheme of the switches SW 1 , SW 2 , SW 3 , SW 4 , SW 5  and SW 6  through the first, second and third states of each of the controllable current generators allow the first and second current sources to operate in an un-switched manner even during time periods where they do not source or sink current to the load circuit. Instead, the superfluous current of a particular current generator is directed through the DC voltage reference  806  by selecting an appropriate setting of the switches. This feature eliminates switching noise for example caused by charge injection from repetitious switching of the first and second current sources when cycling through the first, second and third states. 
     As discussed above in relation to  FIGS. 1-9B , the feedback path circuit  108  can generate an analog feedback signal  124 , which can attenuate or suppress low frequency components in the electrical signal  122  generated by the transducer  101 . 
     Referring again to  FIG. 1A , the second processing circuit  104  receives the low pass filter signal  126  and processes the low pass filter signal  126  to generate a low frequency pressure signal  128 . As discussed above, the second processing circuit  104  includes a compensation circuit  112  that can process the low pass filter signal  126  and generate the low frequency pressure signal  128 . In particular, the compensation circuit  112  can compensate for a non-linear relationships between a sensitivity of the transducer  101  and the change in frequency as well as changes in temperature. 
       FIG. 10  shows an example graph  1000  of sensitivity of the transducer at two different temperatures. In particular, a first plot  1002  depicts the sensitivity over a frequency range at about 10 degrees Celsius, while a second plot  1004  depicts the sensitivity over the same frequency range at about 20 degrees Celsius. Both the first and second plots  1002  and  1004  show that the sensitivity of the transducer varies non-linearly based on the frequency of the incident pressure change. For example, the sensitivity at 10 Hz is about 0 dB, and about −6 dB at about 0.5 Hz. The sensitivity also changes with the change in temperature, due to, for example, changes in the mechanical stresses imposed on the transducer package as a result of changes in temperature. For example, with the change in temperature from about 10 degrees Celsius to about 20 degrees Celsius, the sensitivity at the same frequency changes from about −9.2 dB to about −9.5 dB. In some embodiments, it is desirable that the sensitivity of the transducer  101  is substantially uniform over an operating frequency range, and an operating temperature range. The compensation circuit  112  can process the low pass filter signal  126  to compensate for the changes in sensitivity over changes in frequency and changes in temperature such that the resulting low frequency pressure signal  128  appears as if it were generated based on a uniform sensitivity transducer. 
     In some embodiments, the compensation circuit  112  can include a multiplier that multiplies the amplitudes of frequency components of the low pass filter signal  126  within frequency ranges at which the transducer  101  exhibits non-uniform sensitivity. For example, referring to the first and second sensitivity plots  1002  and  1004  shown in  FIG. 10 , the multiplier can multiply the amplitudes of frequency components of the low pass filter signal  126  that lie in the frequency range where the sensitivity is below 0 dB with appropriate multiplying factors. In some embodiments, the multiplier can determine the appropriate multiplying factor based on a look up table that includes a first column including frequencies, and a corresponding value in another column specifying the multiplying factor. In some other embodiments, a polynomial representing the sensitivity plot of the transducer  101  can be stored in memory, and can be used to determine the multiplying factor. The compensation circuit  112  can similarly compensate for changes in sensitivity with changes in temperature. For example, as shown in  FIG. 1A , the microphone device  100  can include a temperature sensor  113  that can measure the temperature of the microphone device  100 . The compensation circuit  112  can include a look-up table that lists various multiplying factors corresponding to various temperature values. Based on the temperature value received from the temperature sensor  113 , the compensation circuit  112  can determine the appropriate multiplying value from the look-up table. In some embodiment, the look-up tables for frequency compensation can be populated during a calibration process. In some embodiments, the compensation circuit  112  can simultaneously carry out frequency and temperature compensation on the low pass filter signal  126  to generate the low pass filter signal  126 . 
     In some embodiments, the second processing circuit  104  can combine the low pass filter signal  126  and the audio signal  127  to generate the low frequency pressure signal  128 . In some embodiments, the second processing circuit  104  can generate the low frequency pressure signal  128  based solely on the audio signal  127  instead of based on the low pass filter signal  126 . In some such embodiments, the second processing circuit  104  can include a loop filter similar to the LF  118  to process the audio signal  127  and generate a low pass filter signal, which, in turn, can be processed by the compensation circuit  112  to compensate for changes in frequency and temperature to generate the low frequency pressure signal  128 . In some embodiments, the second processing circuit  104  can combine the low pass filter signal  126  and the audio signal  127  to generate the low frequency pressure signal  128  that includes frequency components of both the low pass filter signal  126  and the audio signal  127 . For example, if the desired frequency range for the low frequency pressure signal  128  is up to 20 Hz, but the low pass filter signal  126  includes frequency components only up to 10 Hz, then the second processing circuit  104  can filter the frequency components from about 10 Hz to about 20 Hz from the audio signal  127 , combine the filtered frequency components with the low pass filter signal  126  and generate the low frequency pressure signal  128 . 
       FIG. 11  shows a second example microphone device  1100  having a command and control circuit  1102 . The command and control circuit  1102  can be configured to receive commands and filter configuration data for one or more filters (such as the LF  118 ) from a host processor. The host processor can be a processor associated with a host device in which the second example microphone device  1100  is deployed. The command and control circuit  1102  can include a data line  1104  and a clock line  1106 . The command and control circuit can include a standardized data communication interface according to various serial data communication protocols, such as, for example, I 2 C, Universal Serial Bus (USB), Universal Asynchronous Receiver-Transmitter (UART), SoundWire, and Serial Peripheral Interface (SPI). The command and control circuit  1102  can be configured to encode the audio signal  127  in accordance with the relevant protocol of the data communication interface and provide the encoded audio signal on the data line  1104 . The command and control circuit  1102  is also configured to receive commands and filter configuration data over the data line  1104 . The clock line  1106  can receive a clock signal provided by the host device for synchronizing the data communication. In some embodiments, the second processing circuit  104  may also include a command and control circuit, similar to the command and control circuit  1102 , to receive commands and configuration data for the compensation circuit  112 . The command and control interface of the second processing circuit can encode the low frequency pressure signal  128  to generate an encoded low frequency pressure signal, and provide the encoded low frequency pressure signal over the associated data line. 
     A person skilled in the art will appreciate that at least one or more components of the first processing circuit  102  and the second processing circuit  104  discussed above in relation to  FIGS. 1A-11  can be implemented using an ASIC, such as a digital signal processor (DSP). In some such embodiments, the DSP can be used to implement the LF  118  as a digital filter, such as, for example, a finite impulse response (FIR) filter, an infinite impulse response (IIR) filter, etc. In some implementations, values of the coefficients of the digital filter can be stored in a memory of the DSP. The values of the coefficients can be preloaded into the DSP or can be received via the data line  1104  shown in  FIG. 11 . In one or more embodiments, the at least one or more components of the first processing circuit  102  and the second processing circuit  104  discussed above in relation to  FIGS. 1A-11  can be implemented using ASICs, such as hard-wired logical gates or field programmable gate arrays (FPGAs) synthesized using high level hardware description language, such as Verilog or VHDL. The coefficients of various components, such as filters, can be stored in memory (such as read-only memory (ROM) or metal ROM), stored by way of one-time programmable fuses burned during production, or loaded in memory via the command and control interface, such as, for example, via the command and control circuit  1102 . 
       FIG. 12  shows an exemplary embodiment of a microphone assembly or system  1200 . In some embodiments, the microphone assembly  1200  can be used to implement the microphone devices discussed above in relation to  FIGS. 1A-11 . The microphone assembly  1200  includes a transducer  1202 , such as a MEMS transducer, configured to convert sensed pressure changes into a corresponding electrical signal. The transducer  1202  may, for example, exhibit a transducer capacitance between 0.5 pF and 10 pF. The transducer  1202  may include first and second mutually charged transducer plates, e.g. a diaphragm  1205  and back plate  1206 , respectively, supplying the electrical signal. The microphone assembly  1200  additionally includes a processing circuit  1222 , which may include a semiconductor die, for example a mixed-signal CMOS semiconductor device integrating the various analog and digital circuits discussed above. The microphone assembly  1200  can also include a temperature sensor  1224  that measures the ambient temperature and provides a temperature value (in terms of a voltage/current or a digital value) to the processing circuit  1222 . Bonding wires  1207  can connect the transducer  1202  to the processing circuit  1222 . The processing circuit  1222  is shaped and sized for mounting on a substrate or carrier element  1211  of the microphone assembly  1200 , where the carrier element likewise supports the transducer  1202 . The microphone assembly  1200  includes a housing lid  1203  mounted onto a peripheral edge of the substrate or carrier element  1211  such that the housing lid  1203  and carrier element  1211  jointly form a microphone housing enclosing and protecting the transducer  1202  and the processing circuit  1222  of the microphone assembly  1200 . The microphone housing may include an inlet or port  1209  projecting through the carrier element  1211 , or through the housing lid  1203  in other embodiments, for conveying pressure changes to the transducer  1202 . 
     In one or more embodiments, the carrier element  1211  can include one or more interconnects outside of the housing lid  1203  to send and receive electrical signals to and from the processing circuit  1222 . For example, the interconnects can implement the data line  1104  and the clock line  1106  shown in  FIG. 11 . 
       FIG. 13  shows a block diagram of a third example microphone device  1300 . The third example microphone device  1300  includes several components that are common with the components of the first and the second example microphone devices  100  and  1100  discussed above in relation to  FIGS. 1 and 11 , and such common components have been labeled with like reference numerals. In contrast with the first and the second example microphone devices  100  and  1100 , the third example microphone device  1300  includes a pressure sensor  1301 . The pressure sensor  1301  can measure pressure changes in a frequency range that is below the frequency range of the transducer  101 . For example, the pressure sensor  1301  can be configured to be sensitive to changes in pressure in the frequency range of about 0 Hz (absolute pressure) to about 1 Hz, where the transducer  101  is configured to be sensitive to changes in pressure in the frequency range of about 1 Hz to about 20 kHz. In some embodiments, the pressure sensor  1301  can be configured to be sensitive to pressure changes at all frequencies below the lowest frequency for which the transducer  101  is configured to be sensitive. In one or more embodiments, the pressure sensor  1301  can be implemented using a piezo resistive pressure sensor or a capacitive pressure sensor. 
     The pressure sensor  1301  can sense the changes in pressure (or the absolute pressure) and generate a pressure sensor electrical signal  1322 , which is provided to the second processing circuit  1304 . The second processing circuit  1304  can include at least one of an amplifier, an analog-to-digital converter, and a buffer to process the pressure sensor electrical signal  1322 . The second processing circuit  1304  can receive the low pass filter signal  126  generated by the loop filter  118 . As discussed above, the low pass filter signal  126  can include low frequency components of the electrical signal generated by the transducer  101 . The low frequency components included in the low pass filter signal  126  include frequency components down to a cut-off frequency of the transducer  101 . For example, as discussed above, if the cut-off frequency of the transducer  101  is about 1 Hz, then the low pass filter signal  126  may include frequency components down to 1 Hz. The second processing circuit  1304  includes a cross-over filter circuit  1305  that combines the low pass filter signal  126  and the pressure sensor electrical signal  1322  (which, for example, may be processed by one of the amplifier, the analog-to-digital converter, or the buffer discussed above) to generate a combined pressure signal  1328 . In particular, the cross-over filter  1305  can combine the frequency components from the pressure sensor electrical signal  1322  and the frequency components from the low pass filter signal  126  to generate the combined pressure signal  1328 , which can now include frequency components down to about 0 Hz. This is in contrast with the low frequency pressure signal  128  discussed above in relation to  FIG. 1A , which includes frequency components down only to the cut-off frequency of the transducer  101 . As a result, the third example microphone device  1300  is capable of sensing pressure changes down to absolute pressure. 
       FIG. 14  shows a block diagram of a cross-over filter circuit  1305 , shown in  FIG. 13 . In particular, the cross-over filter  1305  can include a first filter  1403 , a second filter  1404 , an amplifier  1405 , and a summer  1406 . The pressure sensor electrical signal  1322  is provided to the first filter  1403  while the low pass filter signal  126  is provided to the second filter  1404 . The first filter  1403  is a low pass filter while the second filter  1404  is a high pass filter. Both the first and the second filters  1403  and  1404  can be configured to have their respective cut-off frequencies be equal to the cut-off frequency of the transducer  101 . For example, if the cut-off frequency of the transducer  101  is set at 1 Hz, then the cut-off frequencies of the first and the second filters  1403  and  1404  can be set at 1 Hz. In some embodiments, the cut-off frequency of the first filter  1403  can be different from the cut-off frequency of the second filter  1404 . The first filter  1403  filters the pressure sensor electrical signal  1322  to pass all frequency components below the cut-off frequency and generate a first filter output signal  1408 . Likewise, the second filter filters the low pass filter signal  126  to pass all frequency components above the cut-off frequency and generate a second filter output signal  1410 . The amplifier  1405  can optionally amplify the first filter output signal  1408  before providing the first filter output signal  1408  to the summer  1406 . While not shown in  FIG. 14 , an amplifier may also be employed to amplify the second filter output signal  1410  before being fed to the summer  1406 . A gain ‘A’ of the amplifier  1405  can be selected such that the average amplitude of the first filter output signal  1408  is substantially equal to the average amplitude of the second filter output signal  1410 . The summer  1406  adds the first filter output signal  1408  and the second filter output signal  1410  to generate the combined pressure signal  1328 . 
     In some embodiments, the second processing circuit  1304  can also include compensation circuits for compensating for non-linearity of each of the transducer  101  and the pressure sensor  1301 . For example, the second processing circuit can include a compensation circuit that compensates the pressure sensor electrical signal  1322  for non-uniform gain of the pressure sensor  1301  over changes in frequency and temperature, in a manner similar to that discussed above in relation to the compensation circuit  112  shown in  FIG. 1A . In some embodiments, pressure sensor electrical signal  1322  and the low pass filter signal  126  can be compensated by their respective compensation circuits before being provided to the first filter  1403  and the second filter  1404 . 
     The third example microphone device  1300  discussed above in relation to  FIGS. 13 and 14  allows for a significant reduction in size of the pressure sensor  1301  compared to other microphone devices. For example, some microphone devices include both an acoustic transducer and a pressure sensor. The acoustic transducer is configured to have a cut-off frequency of about 20 Hz, or about the lower end of the audible frequency range. The pressure sensor in such microphone devices is used to measure pressure changes in the frequency range of about 0 Hz to about 20 Hz. For such a pressure sensor to have high resolution over the entire frequency range, the pressure sensor has a large size, in the order of about 1 sq. mm. In contrast, the third example microphone device  1300  uses a pressure sensor  1301  that needs to measure a frequency range of only up to about 1 Hz. The remainder of the frequency range, between 1 Hz and 20 Hz, is instead measured by the transducer  101 , which is also used to sense pressure changes in the audible frequency range. This reduction in the range of frequency to be measured by the pressure sensor  1301  results in a pressure sensor  1301  configured with relatively lower resolution. This, in turn, results in a reduction in the size needed for the pressure sensor  1301 . In some embodiments, the pressure sensor  1301  can have a size of only about 0.1 sq. mm. A reduction in size of the pressure sensor  1301  can allow a reduction in the size of the package that houses the third example microphone device  1300 . 
       FIG. 15  an exemplary embodiment of a microphone assembly or system  1500 . The microphone assembly  1500  shown in  FIG. 15  is similar in many respects to the microphone assembly  1200  discussed above in relation to  FIG. 12 . To that extent, like components have been referred to with like reference numbers. The microphone assembly  1500  also includes a pressure sensor  1501  for measuring pressure changes at very low frequencies. For example, the pressure sensor  1501  can be used to implement the pressure sensor  1301  discussed above in relation to  FIG. 13 . For example, the pressure sensor  1501  can be configured to be sensitive to changes in pressure in the frequency range of about 0 Hz to about 1 Hz. In some embodiments, the pressure sensor  1501  can be configured to be sensitive to changes in pressure in the frequency range that is below the frequency range for which the transducer  1202  is configured to be sensitive. 
     In some embodiments, the diaphragm  1205  of the transducer  1202  can include a vent  1503  that allows passage of air between the back volume  1210  and the outside of the microphone assembly  1500 . The size (e.g., diameter or surface area) of the vent  1503  can be selected to set a cut-off frequency of the transducer  1202 . For example, the size of the vent  1503  can be selected such that the cut-off frequency of the transducer  1202  is about 1 Hz. This means that the transducer  1202  is sensitive primarily to the changes in pressure that have a frequency above 1 Hz. Frequencies below 1 Hz can pass through the diaphragm  1205  and be incident on the pressure sensor  1501 . The vent  1503  sets a cut-off frequency, where frequencies above the cut-off frequency are sensed by the transducer  1202  and the frequencies below 1 Hz are sensed by the pressure sensor  1501 . In some such embodiments, the vent  1503  can be seen as an electrical circuit acoustic filter, and in particular as acoustically implementing a cross-over filter (such as the electronic cross-over filter  1305  shown in  FIGS. 13 and 14 ), and the second processing circuit  1304  may not need to implement the cross-over filter  1305 . The size of the vent can be configured to set the cut-off frequency at frequencies other than 1 Hz, such as for example, at frequencies between about 0.1 Hz to about 1 Hz. In some embodiments, the diaphragm  1205  may not have a vent, thereby impeding the pressure changes from being incident on the pressure sensor  1501 . In some such embodiments, the pressure sensor  1501  may be positioned between the port  1209  and the diaphragm  1205 , so that the pressure sensor  1501  can faithfully measure any changes in pressure. In some embodiments, the transducer  1202  can include an electronic filter that electronically sets the frequency of the transducer  1202  to the desired cut-off frequency. In some such embodiments, the electronic filter can include an analog filter, such as a first order or second order analog filter formed using passive components (such as for example, resistors and capacitors), active components (such as for example, transistors and operational amplifiers) or a combination thereof. 
       FIG. 16  shows a flow diagram of a process  1600  for generating a low frequency pressure signal and an audio signal in a microphone device. The process  1600  can be executed by one or more circuits and/or signal processors included in a microphone device, such as the microphone device shown in  FIGS. 1A, 11, and 13 . 
     The process  1600  includes receiving an electrical signal from a transducer, the electrical signal including frequency components in a first frequency range and frequency components in a second frequency range higher than the first frequency range (stage  1602 ). At least one example of this process stage has been discussed above in relation to  FIGS. 1A, 11, and 13 . For example, as shown in  FIGS. 1A, 11, and 13 , the first processing circuit  102  receives an electrical signal  122  generated by the transducer  101 . As shown in  FIG. 1B , the electrical signal can include frequency components in the first frequency range and frequency components in the second frequency range. The first frequency range can include frequencies, for example, between about 0 Hz to about 20 Hz, and the second frequency range can include frequencies, for example, between about 20 Hz to about 20 kHz. 
     The process  1600  includes attenuating, using a feedback signal, frequency components in the first frequency range from the electrical signal to generate an audio signal (stage  1604 ). As discussed above in relation to  FIGS. 1A, 11, and 13 , the feedback signal  124  can be used to suppress or attenuate frequency components in the first frequency range from the electrical signal  122  to generate an audio signal. The feedback signal  124  can be provided by the feedback path circuit  108  and can include frequency components that correspond to the frequency components in the first frequency range included in the electrical signal  122 . 
     The process  1600  includes generating a low pass filter signal by low pass filtering the audio signal where a cut-off frequency of the low pass filter is within the first frequency range (stage  1606 ). As discussed above in relation to  FIGS. 1A, 11, and 13 , the feedback path circuit  108  can provide the audio signal  127  to a LF  118 , where the LF  118  can have a cut-off frequency that is equal to a highest frequency in the first frequency range. The LF  118  can generate a low pass filter signal  126  that includes the frequency components in the first frequency range that appear in the audio signal  127 . In some embodiments, the characteristics of the LF  118 , such as the cut-off frequency, the roll-off rate, the attenuation, etc., can be programmable by programming the values of the filter coefficients of the LF  118 . As discussed in relation to  FIGS. 11 and 13 , the appropriate values of the filter coefficients of the LF  118  can be received via the command and control circuit  1102 . 
     The process  1600  also includes generating the feedback signal based on the low pass filter signal (stage  1608 ). As discussed above in relation to  FIG. 2 , the feedback path circuit includes a DAC  120  that converts the low pass filter signal  126  into a feedback signal  124 , which is provided to the summing node  110 . The DAC  120  can convert the digital low pass filter signal  126  into an analog feedback signal  124 , which can be used to attenuate frequency components in the first frequency range from the electrical signal generated by the transducer  101 . The generation of the feedback signal  124  and the attenuation of the frequency components in the first frequency range from the electrical signal generated by the transducer  101  is discussed above in relation to  FIGS. 2-9B . 
     The process  1600  also includes generating a low frequency pressure signal based on the low pass filter signal (stage  1610 ). As discussed above in relation to  FIGS. 1A, 10, 11 , and  13 , the second processing circuit  104  can include a compensation circuit  112  that can process the low pass filter signal  126  to generate the low frequency pressure signal  128 . The low frequency pressure signal  128  can include frequency components in the first frequency range. 
     The process  1600  may also include generating the low frequency pressure signal based on a pressure sensor signal generated by a pressure sensor included in the microphone device in addition to the transducer. As discussed above in relation to  FIGS. 13-15 , the second processing circuit  1304  can include a cross-over filter  1305  that can combine the pressure sensor electrical signal  1322  generated by the pressure sensor  1301  with the low pass filter signal  126  to generate a combined pressure signal  1328 . 
     The foregoing description of illustrative embodiments has been presented for purposes of illustration and of description. It is not intended to be exhaustive or limiting with respect to the precise form disclosed, and modifications and variations are possible in light of the above teachings or may be acquired from practice of the disclosed embodiments. It is intended that the scope of the invention be defined by the claims appended hereto and their equivalents.