Patent Publication Number: US-7710342-B2

Title: Crossed-dipole antenna for low-loss IBOC transmission from a common radiator apparatus and method

Description:
FIELD OF THE INVENTION 
   The present invention relates generally to radio frequency (RF) electromagnetic signal antennas. More particularly, the present invention relates to dual-feed crossed-dipole circularly polarized broadband antennas for in-band, on-channel broadcasting. 
   BACKGROUND OF THE INVENTION 
   iBiquity Corporation has developed a specification for its “in-band on-channel” (IBOC®) broadcasting system that meets the requirements of the Federal Communications Commission (FCC). Transmitting a hybrid (both analog and digital) IBOC®-compatible broadcast requires radiating an analog signal with frequency modulation (FM) technology and a digital signal with orthogonal frequency division multiplexing (OFDM) technology. The OFDM signal occupies the edges of the FM signal&#39;s emissions mask and has a total radiated power one hundredth (−20 dB) that of the FM signal. Each hybrid IBOC® signal uses one of the hundred radiotelephone channels for public reception established between television channels  6  and  7  in the very high frequency (VHF) band (88.1 MHz to 107.9 MHz). IBOC® also defines standards for all-digital VHF and for AM-band (535 KHz to 1705 KHz) radio. 
   A previous IBOC® antenna design disclosed in U.S. Pat. No. 7,084,822 (“the &#39;822 patent”), incorporated herein by reference, includes crossed dipoles for radiation of analog and digital signals. The propagation concept disclosed includes, in at least one embodiment, two pairs of dipoles in each bay, with the dipoles in each pair spaced horizontally by a quarter wavelength, oriented at right angles to each other within parallel planes, and driven with two substantially unrelated signals, where the two signals are fed as traveling waves from opposite ends of a coaxial line and coupled therefrom to drive the dipoles. 
   A crossed-dipole pair so driven reinforces signal emission at some azimuths and cancels signal emission at other azimuths to produce generally peanut-shaped and overlaid circularly polarized patterns—beams—for the two signals. Each beam has two lobes; the lobes for that beam have the same circular polarization, but are opposite in phase at each instant. The &#39;822 patent discloses a second dipole pair that taps the coaxial line a quarter wavelength from a first dipole pair for impedance cancellation, and that has an azimuthal orientation at right angles to that of the first pair, so that each bay radiates two circularly polarized signals with opposite handedness and oppositely rotating phase. The signals generally fill in at intermediate azimuths to an extent sufficient for the antenna to be termed omnidirectional. 
   While effective, this embodiment is somewhat constrained by the traveling-wave feed method, and is better suited to tower-top mounting and a small number of bays. A second embodiment in the &#39;822 patent feeds crossed dipole pairs from taps on a traveling wave coaxial line, splitting the tapped signals to drive the pairs. This allows all of the radiating elements to be placed to one side of the coaxial line, but is still further limited in power by halving the number of coupling taps per radiator. 
   Another previous IBOC® antenna design is disclosed in copending U.S. application Ser. No. 11/698,065, filed Jan. 26, 2007, titled “Antenna System and Method to Transmit Cross-Polarized Signals from a Common Radiator with Low Mutual Coupling,” incorporated herein by reference. This design includes separate corporate feed from analog and digital transmitters to a plurality of hybrid couplers per bay, each hybrid including unbalanced inputs and balanced outputs, so that multiple crossed-dipole radiators with integral cross-coupling cancellation can be provided in a plurality of bays with low mutual coupling. While highly effective, broad banded (&gt;20% BW for VSWR&lt;1.05:1), and high power capable, this design can be complex, preferably using either a tower-top mounting scheme or a plurality of discrete mountings around a tower or other structure to realize omnidirectional coverage. 
   Multiple-channel broadcast towers are costly to build and occupy significant amounts of real estate in rare locations (high up and near the center of population regions but low in local population, so transmitters can be clustered around them). Many such broadcast towers are relatively full, that is, they are limited in the number of antennas that can be mounted on them with adequate vertical separation, and desirable positions such as tower tops are typically already taken, leaving small or low positions or replacement of existing antennas as enhancement possibilities. Some IBOC®-compatible antenna designs are not readily adapted to tower-side mounting, because they use highly symmetrical structures to achieve omnidirectional patterns and would require robust, extended—and massive—cantilever brackets for tower side mounting. 
   SUMMARY OF THE INVENTION 
   The foregoing disadvantages are overcome, to a great extent, by the present invention, wherein in one aspect a circularly polarized, corporate-feed IBOC®-compliant antenna is provided that in some embodiments affords simplicity in mechanical construction, moderate power capability, high gain, broad bandwidth, good azimuth coverage, adaptability for vertical null, beam tilt, and null fill, little phase runout, and suitability to tower side mounting. 
   In accordance with one embodiment of the present invention, an antenna system for broadcasting radio frequency (RF) electromagnetic (EM) signals over a frequency range is presented. The antenna includes a first pair of crossed dipoles, a second pair of crossed dipoles, a hybrid coupler that includes a first input port, a second input port, a first output port, and a second output port, a first coaxial interconnecting tee from the hybrid coupler first output port to the respective ones of the first pair of crossed dipoles, and a second coaxial interconnecting tee from the hybrid coupler second output port to the respective ones of the second pair of crossed dipoles. 
   In accordance with another embodiment of the present invention, an antenna system for broadcasting radio frequency (RF) electromagnetic (EM) signals, operational over a frequency range, is presented. The antenna includes radiators for radiating an analog frequency-modulated (FM) broadcast-level electromagnetic signal assigned to a channel within the Federal Communications Commission (FCC)-assigned very high frequency public radiotelephone band (VHF band) having a circular polarization, a direction of phase rotation, and a specified extent of gain with respect to a single dipole, and radiators for radiating a digital orthogonal frequency division multiplexed (OFDM) broadcast-level electromagnetic signal assigned to the same channel as the analog signal, having the same circular polarization as the analog signal, opposite direction of phase rotation from the FM signal, and gain that is substantially equal to the gain of the FM signal. In the antenna, the relative power levels of the FM and OFDM signals comply with FCC requirements and further comply with specifications defined by iBiquity® Corporation for In-Band On-Channel (IBOC®) transmission, the radiators for radiating the FM and OFDM signals are positioned at four discrete locations uniformly distributed on a quarter-wavelength square in each of a plurality of vertically-displaced bays, the radiators for radiating the FM signals and the radiators for radiating the OFDM signals are the same physical devices, the FM and OFDM signals are presented to the radiators using corporate feed, and interbay spacing is a function of vertical beam null. 
   In accordance with still another embodiment of the present invention, a method of broadcasting radio frequency (RF) electromagnetic (EM) signals, operational over a frequency range, is presented. The method may include generating a first broadcast signal, generating a second broadcast signal, applying the first signal to a first power divider, applying the second signal to a second power divider, applying a first output signal from the first divider to a first input port of a first 3 dB quarter-wave hybrid coupler, applying a first output signal from the second divider to a second input port of the first hybrid, dividing a first output signal from the first hybrid with a first tee divider, and dividing a second output signal from the first hybrid with a second tee divider. The method may further include applying respective outputs from the first tee divider to a first two orthogonally crossed dipoles, separated by a quarter wavelength, located in parallel planes perpendicular to a ground plane, wherein a line connecting the first-dipole midpoints is orthogonal to the parallel planes of the first two crossed dipoles, and applying respective outputs from the second tee divider to a second two orthogonally crossed dipoles, separated by a quarter wavelength, located in parallel planes perpendicular the planes of the first two dipoles and to a ground plane, wherein a line connecting the second-dipole midpoints is orthogonal to the parallel planes of the second two crossed dipoles. 
   There have thus been outlined, rather broadly, features of the invention, in order that the detailed description thereof that follows may be better understood, and in order that the present contribution to the art may be better appreciated. There are, of course, additional features of the invention that will be described below and which will form the subject matter of the claims appended hereto. 
   In this respect, before explaining at least one embodiment of the invention in detail, it is to be understood that the invention is not limited in its application to the details of construction and to the arrangements of the components set forth in the following description or illustrated in the drawings. The invention is capable of other embodiments, and of being practiced and carried out in various ways. It is also to be understood that the phraseology and terminology employed herein, as well as the abstract, are for the purpose of description, and should not be regarded as limiting. 
   As such, those skilled in the art will appreciate that the conception upon which this disclosure is based may readily be utilized as a basis for the designing of other structures, methods, and systems for carrying out the several purposes of the present invention. It is important, therefore, that the claims be regarded as including such equivalent constructions insofar as they do not depart from the spirit and scope of the present invention. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a perspective view of a multiple-bay antenna according to one embodiment of the instant invention. 
       FIG. 2  is a perspective view of one bay of an antenna according to one embodiment of the instant invention. 
       FIG. 3  is a schematic partial section view of a dipole feed arrangement according to one embodiment of the instant invention. 
       FIG. 4  is a schematic representation of a hybrid coupler illustrating the concepts employed in the instant invention. 
       FIG. 5  is a bottom view of the bay of  FIG. 2 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The invention will now be described with reference to the drawing figures, in which like reference numerals refer to like parts throughout. The present invention provides an apparatus and method that in some embodiments provides a dual-port antenna that supports two isolated broadcasts with substantially null-free, circularly-polarized, rotating-phase propagation patterns, selectable gain, and moderate power handling capability. 
     FIG. 1  shows a multiple-bay crossed dipole antenna  10  in schematic form according to one embodiment of the instant invention. The antenna  10  complies with Federal Communications Commission (FCC) requirements for analog frequency-modulated (FM) broadcast-level electromagnetic signal generation for very high frequency public radiotelephone band (VHF band) broadcasting, and with specifications defined by iBiquity® Corporation for a digital orthogonal frequency division multiplexed (OFDM) broadcast-level electromagnetic signal for In-Band On-Channel (IBOC®) transmission. The antenna  10  uses one hybrid  12  and two pairs of crossed dipoles  14  per bay  16 . A high-power divider  18  for corporate feed of the analog signal and a low-power divider  20  for corporate feed of the digital signal are located within the aperture of the antenna in the embodiment shown. For other embodiments, the dividers  18  and  20  may be fitted at any suitable location, such as at a tower base (not shown). Such factors as wind and weight loading of the dividers  18  and  20  may be offset by wind and weight loads of individual coaxial lines  102  coupling the dividers  18  and  20  to the hybrids  12  in some of these embodiments. 
   Feed lines  102  from the dividers  18  and  20  to the individual hybrids  12  in the bays  16  are equal in length in a realization of the embodiment shown. This configuration, in conjunction with providing dividers  18  and  20  that are substantially uniform in transit time from an input port to all output ports, can provide low phase runout, wherein phase runout is a factor degrading beam precision. In other embodiments, closer-in feed lines  102  can be made shorter by, for example, a wavelength per bay  16 ; this may reduce weight and wind loading while reducing performance to some extent. Other embodiments, such as ones which may use traveling wave feed lines in lieu of a power divider, may feed successive bays with successively delayed signals, increasing phase runout in exchange for structural robustness and configuration simplicity. 
   Signals for the antenna of  FIG. 1  originate in an analog transmitter  104  and a digital transmitter  106 , shown schematically, with at least the digital transmitter  106  protected by a circulator  108  and a dissipative load  110 , connected by respective coaxial lines  112  and  114  from a location for the transmitters  104 ,  106  that is near an antenna tower (not shown) in at least some embodiments. Electrical power, broadcast information sources, connections thereof to the respective transmitters  104 ,  106 , station loads, a tower, and other apparatus required for a complete broadcasting facility, are not shown in  FIG. 1 . 
   The antenna of  FIG. 1  provides a plurality of bays  16  of the form of  FIGS. 2 and 5 , with gain realized by spacing the bays  16  at preferred vertical intervals  116  and by aligning dipoles having corresponding azimuth orientations in the respective bays  16  so that synchronous rotating-phase signals are emitted from all bays  16 . An antenna having a single bay  16  of the configuration shown may be preferred in some embodiments. 
   Bandwidth in the embodiment shown may be widened by combining large element diameter, selection of connector, hybrid, and power divider designs, providing short, low-loss, and/or equal-length coaxial lines, and the like. Multiple low-level- or high-level-combined channels may be present in each of the transmitter apparatuses  104  and  106  shown. 
     FIG. 2  shows a single bay  16  of an antenna  10  shown in  FIG. 1 . A single hybrid  12  within the bay  16  shown has a high-power coaxial (unbalanced) input fitting  24 , having an outer-conductor flange  26  and an inner conductor coupling  28 , known in the art as a “bullet”, and a low power coaxial (unbalanced) input fitting  30 , having an outer conductor mounting flange  32  and an inner conductor bullet  34 . The hybrid  12  has two coaxial output lines  36  and  38 , respectively, terminating in coaxial crossbars  40  and  42  that divide the signals applied to them into substantially equal portions. The portions in the first indicated crossbar  40  propagate outwardly with equal phase to excite terminal dipoles  44  and  46 , while the portions in the crossbar  42  propagate outwardly with equal phase to excite terminal dipoles  48  and  50 . 
   Junction impedance between the hybrid output lines  36  and  38  and the respective coaxial crossbars  40  and  42 —each representing two loads in parallel—can be matched by doubling the relative line impedance of the latter. 
                 Z   =     K   ⁢       log   ⁢           ⁢     (     D   d     )         ɛ                 (   1   )               
where
 
   Z=impedance 
   K=a proportionality constant 
   D=outer conductor inner diameter 
   d=inner conductor outer diameter 
   ∈=dielectric constant (epsilon) 
   For example, by decreasing the crossbar inner conductor diameter d or by filling the output lines  36  and  38  with an insulator having a relatively high dielectric constant while leaving the crossbars  40  and  42  air-filled, impedance can be readily matched. Other impedance-matching methods are also well known in the art, and the foregoing methods should not be viewed as limiting. Flanges  56  shown at the entrances to the crossbars  40  and  42  and to the dipoles  44 ,  46 ,  48 , and  50  are commonly employed for convenience in manufacture, and likewise should not be viewed as limiting. Radiused dipole ends  58  as shown are one of several known approaches for controlling electrostatic discharge, bandwidth, and other properties, and should likewise not be viewed as limiting. 
     FIG. 3  shows, in section, a largely schematic arrangement  60  for coupling an inner conductor  62  of a crossbar to a “hot” monopole  64  of a terminal dipole  66 . The crossbar outer conductor  68  has electrical continuity with this terminal dipole&#39;s “cold” monopole  70 , while the crossbar inner conductor  62  feeds past an insulating section  72  to connect to the hot monopole  64 . A joining location  74  includes a conductive wafer  76  brazed or otherwise electrically coupled to the hot monopole  64  near the cold monopole  70 . This is one of several well-known joining methods, each typically having particular impedance and propagation characteristics, and should not be viewed as limiting. Methods for fine adjustment of dipole length and for sealing the interior volume of the antenna against contaminants are well known in the art, are not critical to the illustrated schema, and are not detailed in the section view of  FIG. 3 . Likewise, internal arrangements for the flanges  56  of  FIG. 2  are not critical to the dipole feed function and are not detailed in this section view. Similarly, insulating inner-conductor positioning spacers  78  are shown largely schematically; in practice, such spacers can have many forms, and are chosen for suitability to a specific embodiment. 
   In some embodiments, center-fed dipoles having lengths approximating a half wavelength may be employed. However, as is well known in the art, performance approaching that of full-size dipoles may be realized by shortening the dipoles and moving and configuring the driving point sufficiently to maintain a preferred value of impedance. While an arrangement of the latter kind applies for the embodiment shown, this should not be viewed as limiting. 
     FIG. 4  schematically illustrates a coaxially-fed hybrid coupler  80 . A first input signal  82 , applied to a first input port  84 , is divided in half (depending on exact dimensions of coupler  80  structure and the frequency of the applied signal), with a first half coupled electromagnetically to a first output port  86  with nominal (reference) delay and a second half conveyed conductively to a second output port  88  with one-quarter wavelength of additional delay. A coupler  80  of proper design and correct first input signal  82  frequency reduces or prevents signal  82  leakage at a second input port  92 . A second input signal  90 , applied to the second input port  92  and treated like the first input signal  82 , provides a reference-delay half emitted at the second output port  88 , a quarter-wave-delayed half emitted at the first output port  86 , and substantially no leakage at the first input port  84 . Isolation between the two input signals  82  and  90  in at least some embodiments can be on the order of 30 dB or better. 
   This so-called 3 dB, 90 degree, or quarter-wave hybrid coupler, combiner, splitter, or divider  80  has many applications in the art. Geometries other than the indicated rectangular stripline are used for this and other frequency ranges, power ratios, and relative phase angles, so that the configuration shown should not be viewed as limiting. For example, a so-called magic tee hybrid produces a 180 degree delay (one-half wavelength) in an open line, coaxial line, stripline, or waveguide realization if configured for a suitable frequency range and power level. Power ratios other than 3 dB (e.g., 6 dB, 10 dB, 20 dB) may be realized by adjusting dimensions and frequencies for a given application. The hybrid shown in  FIGS. 1 ,  2 , and  5  has a horseshoe-shaped internal layout that allows placement of inputs and outputs in the spatial locations indicated in those figures while realizing 3 dB power split, quarter-wave phase shift, and isolation between input ports for a specified frequency range. Other hybrid configurations may provide comparable capability, and may be preferred in some embodiments. 
   Returning to  FIG. 4 , a correctly configured hybrid, as discussed above, effectively isolates two applied signals  82 ,  90  from each other, including masking the digital (OFDM) input port ( 92  in  FIG. 4 , at  30  in  FIG. 2 ) from the analog (FM) input port ( 84  in  FIG. 4 , at  24  in  FIG. 2 ), so that the high-power signal  82  applied to the analog port  84  is substantially prevented from propagating to the digital transmitter ( 106  in  FIG. 1 ). As a result, reduced protection is needed to prevent the digital transmitter  106  from being overloaded or modulated by the analog transmitter ( 104  in  FIG. 1 ), while the analog transmitter  104  is substantially immune from overload or modulation by the digital transmitter  106  because of both this isolation and the greater output power of the analog transmitter  106 . Thus, in a typical embodiment, a circulator ( 108  in  FIG. 1 ) and dissipative load ( 110  in  FIG. 1 ) of modest power handling capability are sufficient to support operation of a properly-sized digital transmitter  106  and a likewise properly-sized analog transmitter  104  for IBOC® applications. 
     FIG. 5  shows a bottom view of the bay  16  shown in  FIG. 2 . The crossbars  40  and  42  are shown at right angles to each other. The angle from the cross bar  40  to the output coaxial line  36  from the hybrid  12  is approximately 45 degrees in this embodiment; this is one of several realizable arrangements, and should not be viewed as limiting. The feed arrangement for the high power hybrid input  24  having a flange  26  and a center conductor bullet  28 , is also shown. All four dipoles  44 ,  46 ,  48 , and  50  are oblique to the viewing plane. 
   It may be properly inferred that the dipoles  44  and  46  are coupled to the center conductor of associated crossbar  40  by an arrangement comparable to that shown in  FIG. 3 . The dipoles  44  and  46  of this pair, separated by one-quarter wavelength, are driven in phase and spatially rotated by 90 degrees to each other with the relative orientation shown in  FIGS. 2 and 5 . As a consequence, a component of a first signal, fed to the high-power port  24 , emitted from a first dipole  44 , propagated in the direction of the second dipole  46 , and reinforced by a corresponding component of the first signal emitted from the second dipole  46 , forms a circularly polarized signal with a particular handedness. For example, assuming that applied signals having horizontal and vertical components E H  and E V , respectively, appear as E 1  at a first dipole  44  rotated 45 degrees from the horizontal in a positive direction, and as E 2  at a second dipole  46  rotated a like amount in a negative direction,
 
 E   1   =E   H1   +E   V1   =E   1  cos θ+ E   1  sin θ  (2)
 
 E   2 =E H2   +E   V2   =E   2  cos(−θ)+ E   2  sin(−θ)=− E   2  cos θ+ E   2  sin θ  (3)
 
   For β=distance between the radiators in wavelengths (λ), the instantaneous sum S of the signals E 1  and E 2  is
 
 S=E   1   +E   2  cos β  (4)
 
Let E 1 =E 2 =E, i.e., equal signals applied in phase to the respective dipoles. Then
 
 E   H   =E  cos θ+(− E  cos θ cos β)= E  cos θ(1−cos β)  (5)
 
 E   V   =E  sin θ+( E  sin θ cos β)= E  sin θ(1+cos β)  (6)
 
If β=π/2 [i.e., 90 degrees, or λ/4], then cos β=cos(π/2)=0. Then
 
E H =E cos θ  (7)
 
and
 
E V =E sin θ  (8)
 
If β=π, then cos β=cos π=−1. Then
 
E H =0  (9)
 
and
 
E V =2E sin θ  (10)
 
   Thus, with dipole spacing of one quarter wavelength, horizontal and vertical components are equal, achieving approximately circular polarization. However, with dipole spacing of one half wavelength, the horizontal component is zero, all of the energy is in the vertical component, and a vertical linear output polarization is realized. Similarly, changing the spacing β to one wavelength realizes horizontal linear output polarization. 
   Signals emitted from each dipole in the direction of the other form lobes having the same handedness of circular polarization. The lobes are opposite in polarity, however—that is, with reference to the midpoint of the crossbar, the lobes differ by 180 degrees in both phase and azimuth. 
   Signal components at azimuths perpendicular to these lobes largely cancel at far field, as the dipoles are oppositely polarized but equal in phase, and emit proximally. Signal energy at intermediate azimuths reinforces to an intermediate extent and retains circular polarization. Unlike some radiator configurations, the crossed dipoles form similar beams in azimuth and elevation, so two circularly-polarized lobes in a peanut pattern are formed. 
   The hybrid  12  delays the first signal to the distal output coax  38  (not shown in  FIG. 5 ) by an additional 90 degrees, so first-signal emission from dipoles  48  and  50  is identical to that of dipoles  44  and  46  but delayed by 90 degrees. Therefore, a signal peak occurs on dipole  44 , followed by a peak on dipole  48 , 90 degrees thereafter, then on dipole  46 , 180 degrees after dipole  44 , and on dipole  50 , 270 degrees after dipole  44 . Thus, not only does the first signal produce a second circularly-polarized peanut lobe pattern on dipoles  48  and  50 , but the four dipoles produce a signal having rotating phase that advances clockwise in azimuth ( 44 - 48 - 46 - 50 ). 
   A second signal, fed to the hybrid  12  at the low-power port  30 , shown in  FIG. 2 , emits first from the distal dipoles  48  and  50 , then the foreground dipoles  44  and  46 . As phased by the hybrid  12 , the second signal is also right-hand circularly polarized, but rotates counterclockwise in azimuth ( 48 - 44 - 50 - 46 ). 
   Vertical placement of bays  16 , shown in  FIG. 1 , may be at any of several intervals. In many embodiments, a user will seek to reduce a number of bays  16  in an available aperture consistent with available transmitter  104 ,  106  power output, thereby reducing material cost, complexity, and wind loading while having comparatively little effect on gain and spurious beam propagation. Some of these embodiments provide vertical-radiation nulls—that is, the embodiments minimize mutual coupling between radiators while avoiding generating strong downward beams and wasted upward beams. The nulls in an elevation pattern with all bays  16  driven in phase are defined by: 
                 δ   =       sin     -   1       ⁡     (       k   ⁢           ⁢   λ     nd     )               (   11   )               
where
 
   δ=null angle 
   k=an integer 
   d=distance between bays 
   n=number of elements 
   λ=wavelength 
   k≠n (this is a critical consideration: whole-number-wavelength spacing does not work.) 
   To minimize downward radiation and interbay coupling, a null at δ=90 degrees is required: 
   
     
       
         
           
             
               
                 
                   
                     
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   It will be noted that the most aperture-efficient spacing is 
                       (     n   -   1     )     ⁢             n     ⁢   λ           (   14   )               
—that is, close to but less than one wavelength. Closer spacings have other drawbacks, such as lower antenna gain in proportion to complexity, and thus higher wind loading and material and operating cost in proportion to broadcast coverage. Wider spacings can lead to grating lobes (side lobes replicating the main beam; see Johnson, R. C.,  Antenna Engineering Handbook,  3 rd Edn ., McGraw-Hill, 1993, pp. 3.7, 3.22, 19.6-7, 20.6) as well as increased tower footprint and reduced efficiency. Thus, for example, if an aperture of four wavelengths of tower height (plus gaps between the antenna in question and those above and below) is available, then n−1=4, the number of radiators is 5, and a spacing of 0.8 wavelengths between adjacent bays is the value that may be preferred for many embodiments.
 
   It is to be understood that other considerations may override this optimization for some embodiments. Beam tilt, for example, may dictate some adjustment to the indicated (uniform) spacing, while null fill may be provided by making the spacing nonuniform, while retaining spacing near (n−1)/n. Spacings other than (n−1)/n may be appropriate for still other embodiments, such as those having abundant transmitter power available, or not requiring a vertical null. At another extreme, a single-bay configuration conforms to the description, with a vertical spacing between bays of zero. 
   Vertical displacement between the crossbars  40  and  42  in the embodiment shown in  FIG. 2  is a small fraction of a wavelength, and is of little net effect. Other feed arrangements, such as ones that place the pairs of dipoles more nearly at a common height or more displaced vertically, are also feasible, so that cost and other secondary considerations may dictate layout within each bay  16 . In all cases, however, the four dipoles  44 ,  48 ,  46 , and  50  of each bay  16  may be seen to be approximately centered on respective edges of a planar square parallel to a ground plane representing the surface above which the antenna is mounted, parallel to an effective radiation plane of the antenna  10 , and intermediate between the coaxial feed lines  36  and  38  directed to the crossbars  40  and  42  of the bay  16 , the perimeter of which square lies in the planes of the respective dipoles  44 ,  48 ,  46 , and  50 . 
   Distance from the hybrid  12  to the crossbars  40  and  42  is not required to be a tuned length. As a consequence, any length may be selected for the coaxial feed lines  36  and  38  from the hybrids  12 , in keeping with structural considerations (ice and wind loading, etc.) and interrelationship between the tower and the achieved radiation pattern. 
   The antenna is made substantially omnidirectional by having relatively equal lobes spaced at 90 degrees in azimuth and limiting nulls between lobes. The lobes are oblique to the feed hybrids  12  in the embodiment shown, so that only slight pattern degradation is caused by mounting the antenna alongside a guyed or freestanding tower. Any metallic or otherwise reflective tower parts may affect the achieved pattern inversely to configuration and distance from the respective tower parts to the antenna dipoles  44 ,  48 ,  46 , and  50 . Orientation may be optimized with known antenna ray tracing software followed by validation testing and adjustment. Installed height and the presence of other antennas on the tower will likewise affect final far-field signal characteristics. 
   The many features and advantages of the invention are apparent from the detailed specification, and, thus, it is intended by the appended claims to cover all such features and advantages of the invention which fall within the true spirit and scope of the invention. Further, since numerous modifications and variations will readily occur to those skilled in the art, it is not desired to limit the invention to the exact construction and operation illustrated and described, and, accordingly, all suitable modifications and equivalents may be resorted to that fall within the scope of the invention.