Patent Publication Number: US-5894334-A

Title: Carrier recovery system for a vestigial sideband signal

Description:
FIELD OF THE INVENTION 
     This invention concerns a digital signal processing system. In particular, the invention concerns a carrier recovery system for use in a receiver of a vestigial side band (VSB) signal such as may be modulated with high definition television (HDTV) information, for example. 
     BACKGROUND OF THE INVENTION 
     The recovery of data from a VSB or a QAM (Quadrature Amplitude Modulated) signal at a receiver requires the implementation of three functions: timing recovery for symbol synchronization, carrier recovery (frequency demodulation) and equalization. Timing recovery is the process by which the receiver clock (timebase) is synchronized to the transmitter clock. This permits the received signal to be sampled at the optimum point in time to reduce the chance of a slicing error associated with decision-directed processing of received symbol values. Carrier recovery is the process by which a received RF signal, after being frequency shifted to a lower intermediate frequency passband, is frequency shifted to baseband to permit recovery of the modulating baseband information. Equalization is a process which compensates for the effects of transmission channel disturbances upon the received signal. More specifically, equalization removes baseband intersymbol interference (ISI) caused by transmission channel disturbances including the low pass filtering effect of the channel. ISI causes the value of a given symbol to be distorted by the values of preceding and following symbols. 
     For QAM signals, timing recovery is usually the first function implemented in a receiver. The timing is recovered from either the intermediate passband signal or from a near-baseband signal, i.e., a baseband signal with a carrier offset that is corrected by a carrier recovery network. In either case, timing can be established prior to baseband demodulation. The carrier recovery demodulation process is usually a two step process. First, the passband signal is demodulated to near-baseband by a frequency shifter which uses a &#34;best guess&#34; as to what the frequency offset is between the incoming passband signal and the desired baseband signal. This frequency shift is usually performed by analog circuits; i.e., prior to analog to digital conversion in the receiver. Next, equalization is performed on this near-baseband signal. Finally, carrier recovery is performed which removes any residual frequency offsets from the near-baseband signal to produce a true baseband output signal. This function is performed by digital receiver circuits. The equalizer is inserted between a first local oscillator which performs the shifting to near-baseband, and the carrier recovery loop network. This is because the carrier recovery process typically is a decision-directed process (as known) that requires at least a partially open &#34;eye&#34; which is provided by the equalizer function. 
     A QAM signal is represented by a two-dimensional data symbol constellation defined by Real and Imaginary axes. In contrast, a VSB signal is represented by a one-dimensional data symbol constellation wherein only one axis contains quantized data to be recovered at a receiver. Synchronous demodulation of a VSB signal has usually been accomplished with the aid of a pilot signal. The pilot signal facilitates demodulating the VSB signal to baseband in one step, typically without residual phase or frequency errors. Performing the functions of timing recovery, demodulation and equalization in the order they are performed for QAM signals does not work for VSB signals using conventional techniques. For QAM signals, several timing recovery methods are known which are independent of the frequency offset between the near-baseband signal and the baseband signal. However, it is generally accepted that frequency independent timing recovery is not feasible for VSB signals. For this reasons, in VSB systems, absolute demodulation to baseband has historically been implemented first. 
     In a pilot-assisted VSB system, the pilot component is injected into the baseband signal at the transmitter as a small DC offset. This DC offset generates a carrier &#34;tone&#34; because when the DC offset is multiplied by an alternating signal such as in the form of a high frequency cosine function, a similarly phased carrier tone results. This carrier tone can be used by a phase locked loop (PLL) in the receiver demodulator to translate the modulated VSB signal to baseband. Since the pilot represents the DC component of the VSB signal, and this DC component is in the middle of the vestigial sideband, the tone appears to be situated in the middle of modulation &#34;noise&#34; caused by the data itself. Normally this modulation noise is treated as unwanted signal for the pilot tracking PLL at the receiver. 
     The pilot can be extracted by means of a very narrow bandpass filter prior to the PLL. However, this requires a filter with such a narrow bandwidth to achieve sufficient data noise rejection that the tracking PLL is not capable of tracking all phase and frequency offsets of the incoming signal, especially when consumer grade tuners are used. Shifting the pilot component to baseband is not done easily in the digital domain. If analog circuits are used, problems of compensation for tolerances and temperature effects, for example, will have to be addressed. The pilot component also wastes power, and it is possible that a transmission channel perturbation such as a channel null will cancel the pilot. For these and other reasons, it is herein recognized as desirable to provide a system capable of demodulating a VSB signal to baseband without relying on a pilot component of the received signal. 
     One example of a VSB system including a pilot component is the Grand Alliance HDTV transmission system recently proposed for the United States. This system employs a VSB digital transmission format for conveying a packetized datastream, and is being evaluated in the United States by the Federal Communications Commission through its Advisory Committee of Advanced Television Service (ACATS). A description of the Grand Alliance HDTV system as submitted to the ACATS Technical Subgroup on Feb. 22, 1994 (draft document) is found in the 1994 Proceedings of the National Association of Broadcasters, 48th Annual Broadcast Engineering Conference Proceedings, Mar. 20-24, 1994. 
     SUMMARY OF THE INVENTION 
     In accordance with the principles of the present invention, a carrier recovery demodulation system suitable for use with VSB signals advantageously achieves carrier recovery without reliance on a pilot signal or an equivalent signal. This is accomplished using a filter network and a phase detector responsive to output signals from the filter network. The filter network has a band edge response with respect to at least one of upper and lower band edges of the VSB signal frequency spectrum, for producing a suppressed carrier double sideband amplitude modulated (AM) signal at the output of the filter network. 
     In accordance with a feature of the invention, the filter network comprises first and second digital band edge filters having band edge responses respectively associated with upper and lower band edges of the VSB signal frequency spectrum for producing double sideband amplitude modulated signals at respective outputs of the first and second filters. The filter responses are complementary to the frequency spectrum of said received VSB signal at the band edge that the first and second filters are respectively filtering. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawing: 
     FIG. 1 is a block diagram of a portion of an advanced television receiver, such as a HDTV receiver, including a carrier recovery system in accordance with the principles of the invention. 
     FIGS. 2-6 illustrate amplitude versus frequency responses of signals associated with the operation of the system of FIG. 1. 
    
    
     DETAILED DESCRIPTION OF THE DRAWINGS 
     In FIG. 1, a broadcast VSB modulated HDTV analog signal received by an antenna 10 is processed by an input network 14 including RF tuning circuits, a double conversion tuner for producing an intermediate frequency passband suitable for conversion to digital form and appropriate gain control circuits, for example. The received VSB signal illustratively is an 8-VSB signal with a symbol rate of approximately 10.76 Msymbols/second occupying a conventional NTSC 6 MHz frequency spectrum, in accordance with the Grand Alliance HDTV specification. Specifically, the received VSB signal in this example is an 8-VSB signal having a one dimensional data constellation defined by the following eight data symbols: 
     
         -7 -5 -3 -1 1 3 5 7 
    
     The Nyquist bandwidth for this system is nominally 5.38 MHz for example, with excess bandwidth of nominally 0.31 MHz at each bandedge. The disclosed system may also be used for 16-VSB signals, for example. 
     The output signal from input processor 14 is converted from analog to digital form by an analog-to-digital converter 16, which operates at a sample rate of 2 samples/symbol. The received VSB signal may include a pilot component and has been demodulated by unit 14 so that the center of the 6 MHz band is nominally situated at 5.38 MHz. The frequency spectrum of this signal at the input of ADC 16 occupies a range of 2.38 MHz to 8.38 MHz. When timing synchronization is established, ADC unit 16 samples this signal at 21.52 MHz, which is twice the symbol rate. The pilot component, which represents the DC point of the original baseband pulse amplitude modulated (PAM) signal, is nominally situated at 2.69 MHz (the Nyquist frequency), which is 1/8 f sr . In the following discussion 
     f c  is the carrier frequency of the transmitted signal (nominally 5.38 MHz), 
     f st  is the transmitted symbol frequency (10.76 Msymbols/sec, or four times the Nyquist frequency), and 
     f sr  is the receiver sampling frequency (21.52 Mhz). At timing lock, f sr  =2f st . At carrier lock when demodulation to baseband results, f c  =1/4 f sr . 
     The digital signal from ADC unit 16 is applied to two complex bandedge filters 20 and 22, which are mirror image filters around the Nyquist frequency. Each filter exhibits real and imaginary functions so that output signals from these filters contain real and imaginary components. In FIG. 1, a letter &#34;C&#34; designates those signal paths which convey complex signals with real and imaginary components. Other signal paths convey only real components. Filters 20 and 22 produce output signals without image components, i.e., the output signals contain either positive or negative spectral components but not both. This has the advantage of not generating spurious components that may be difficult to remove subsequently. In this system filters 20 and 22 are designed to have complex analytic output signals with negative spectral components as will be seen from FIG. 2. The negative spectrum is arbitrary; the positive spectrum also could have been chosen. 
     FIG. 2 depicts the negative frequency spectrum encompassed by the bandpass responses of filters 20 and 22, and by the bandwidth of the received VSB signal as applied to the inputs of filters 20 and 22. The input real signal has positive and negative spectra. The positive spectrum is cancelled using known techniques, leaving the negative spectrum. Filter 20 extracts the upper bandedge of the negative spectrum of the VSB signal, and filter 22 extracts the lower bandedge of the negative spectrum of the VSB signal. The upper bandedge is that which contains the highest frequency components, regardless of whether they are positive or negative components. The lower bandedge is associated with the lowest frequency components. The bandedge responses of filters 20, 22 and the VSB signal intersect at the Nyquist points. In FIG. 2 and subsequent Figures, a symbol &#34;Δ&#34; designates a carrier frequency offset such as may be associated with a near-baseband signal, i.e., a signal not completely frequency shifted to baseband. This offset will be discussed in detail in connection with the carrier recovery (baseband demodulation) network. 
     The responses of filters 20 and 22 are complementary to the frequency spectrum of the input signal at the bandedge the filters are extracting, as shown in FIG. 2. This has the effect of producing a double sideband suppressed carrier amplitude modulated (AM) signal when no pilot component is present in the received VSB signal (FIG. 3), and producing a double sideband residual carrier AM signal when a pilot is present in that bandedge. The response of filter 20 to the left of frequency f 1  is not critical, and the response of filter 22 to the right of frequency f 2  is not critical. 
     Prior to establishing timing and carrier lock, these AM signals contain frequency (and phase) offsets that may be used for timing and carrier recovery. Specifically, the center of the AM signal obtained from the output of upper bandedge filter 20 is situated at -f c  -1/4 f st . If a pilot signal is present (such as in the Grand Alliance HDTV system), it would appear at this frequency. The frequency 1/4 f st  is one-fourth the symbol frequency if the signal is treated as a VSB signal. Similarly, the center of the AM signal obtained from the output of the lower bandedge filter is situated at -f c  +1/4 f st  (the Nyquist frequency). Timing synchronization is achieved when the frequency of the sampling clock input (CLK) of ADC unit 16 is four times the frequency difference between the carriers of these two upper and lower suppressed carrier band edge AM signals (FIG. 3). 
     The timing recovery system operates as follows. The output signal from filter 22 is conjugated by unit 25 to flip the spectrum of the filter 22 output signal from negative to positive. This is illustrated by FIG. 4. Conjugation is a well-known process performed in unit 25 by first separating the signal into its real and imaginary components using well-known methods. The imaginary component is inverted by multiplying it by a negative unity factor. The inverted imaginary component and the original real component are re-combined. The recombined lower bandedge AM signal is multiplied with the upper bandedge AM signal from filter 20 in multiplier 26. An AM signal produced at the output of multiplier 26 has carrier frequency component f c  removed, as illustrated by FIG. 5. Removal of this f c  component results because the negative carrier of the upper bandedge component cancels the positive carrier of the conjugated AM signal from the lower bandedge. The AM signal at a suppressed carrier center frequency of 1/2 f st  is maintained because both of the multiplied AM signals are double sideband signals. In the frequency domain these signals are represented at baseband as even functions (i.e., as real only components), and their convolution is represented at baseband as even valued functions. 
     The bandwidth of the AM output signal from multiplier 26 has been doubled by the convolution process (multiplication in time produces convolution in frequency). By driving every other sample to zero, the frequency of the receiver sampling clock CLK can be synchronized to the symbol frequency of the input VSB signal independent of any carrier offset (Δ). This is accomplished by a phase detector 28 in a timing recovery network 30 as follows. 
     The imaginary component of the double sideband AM output signal from multiplier 26 (FIG. 5) is an indication of the magnitude of signal mis-timing. The real component indicates the direction of the mis-timing (AM suppressed carrier signals have an ambiguity of 180 degrees which is to be resolved). If this AM signal is perfectly timed, the imaginary component is absent. The double sideband AM signal from multiplier 26 is separated into its constituent real and imaginary components by means of a unit 32 in phase detector 28 using known separation techniques. Using known techniques, a translation unit 34 determines the sign of real component (for direction information), and multiplies this sign by the separated imaginary component samples. The output of multiplier 36 represents an error signal which is driven to zero by the action of the timing control loop when timing lock is achieved. 
     Since the carrier frequency of the double sideband signal is nominally situated at 1/2 f sr , at lock of the imaginary component of the output signal from multiplier 36 will be zero. Multiplying the imaginary component with the sign of the real component gives phase detector 28 the ability to differentiate between positive and negative frequency offsets. 
     The output signal from phase detector 28 is filtered by a low pass loop filter 38 which contains both an integral path and a proportional path, as known, and is clocked at a frequency of 1/2 f sr . Loop filter 38 is clocked to process every other sample of the input signal since the purpose of the loop is to drive every other sample of the imaginary component to zero. The output of filter 38 is a DC voltage which is applied to a voltage controlled oscillator (VCO) 40. Oscillator 40 provides the receiver sampling clock CLK for ADC unit 16 as a function of the DC voltage. Timing synchronism is achieved when the ADC sampling clock provided by the described timing recovery system including network 30 is four times the frequency difference between the carriers of the two AM output signals from filters 20 and 22 (FIG. 3). The proportional and integral control portions of filter 38 are adjustable as known by using K1 and K2 gain control scalars respectively. These scalars are set to large values to facilitate signal acquisition in the acquisition mode, and may be reduced in value during the tracking mode to increase noise immunity. The time required to achieve timing lock varies as a function of the amount of noise and multipath distortion present in the signal, the control loop bandwidth, and the control loop time constants, for example. 
     The carrier can be recovered using two different methods in the system of FIG. 1. One method uses both of the bandedge AM signals from the outputs of filters 20 and 22 in a manner that is similar to that described above for timing recovery. The second method uses only one bandedge of the received signal. In the second case, typically the bandedge which is used is that which may contain the pilot. The extra energy associated with the pilot component enhances the performance of the carrier recovery loop in low signal-to-noise conditions. It is noted, however, that both of these methods advantageously do not require the presence of a pilot component. 
     The carrier recovery method using both band edges multiplies the outputs of filters 20 and 22 together in a multiplier 45 without conjugating the signal as was done for timing recovery. This multiplication produces a suppressed carrier AM signal at the output of multiplier 45 with a carrier frequency of -2f c . The symbol rate component f st  has been completely removed from this AM signal. If a carrier offset (Δ) exists, the carrier frequency is at -2f c  -2Δ as depicted in FIG. 6. Up to this point, the carrier recovery process is independent of the receiver demodulator sampling clock frequency, f sr . 
     In digital signal processing applications, it is generally desirable to design voltage controlled oscillators (VCOs) or spectral shifters that conveniently produce signals which are harmonically related to the clock frequency of the digital signal processor. In this regard it is noted that the complex double sideband AM signal at the output of multiplier 45 (FIG. 6) is centered at a frequency of -2f c  (neglecting any carrier offset), or at -2fc-2Δ (including the offset). This means that this AM signal is straddling the aliasing foldover region. More specifically, in practice, the left sideband portion of the AM signal shown in FIG. 6 actually &#34;wraps around&#34; into the positive frequency spectrum. Aliasing does not occur because this AM signal is a complex signal in which the adjacent positive frequency component of the first negative repeat band has been removed. 
     To simplify the design of the carrier recovery network, an associated phase detector 54 is essentially the same as phase detector 28 used in timing recovery network 30. In order to do achieve this, the carrier of the AM input signal to phase detector 54 must be driven to 1/4 f sr . The carrier of the AM signal input to phase detector 54 is nominally shifted to a frequency of 1/4 f sr  by using a complex spectral shifter operating at +1/4 f sr . The spectral shifter comprises complex multipliers 52 and 64. Multiplier 64 responds to a 1/4 f sr  sampling signal for shifting the output signal of VCO 62 by +1/4 f sr . The response of VCO 62 is similar to VCO 40 in the timing control loop, and responds to a DC voltage produced by a low pass loop filter 60 which is similar to filter 38 in the timing recovery loop. A resulting complex output signal from multiplier 64, which contains the frequency offset generated by the loop plus the fixed 1/4 f sr , is applied to an input of complex multiplier 52. The other input of multiplier 52 receives the AM signal centered at -2f c  from the output of complex multiplier 45. 
     Phase detector 54 operates the same as phase detector 28 in the timing recovery loop. Phase detector 54 includes a real/imaginary component separator 55, a sign function translation network 56, and an output multiplier 57. Phase detector 54 operates by multiplying the imaginary component of the output signal from multiplier 52 with the sign of the real component. This causes the value of every other sample of the imaginary component to be driven to zero. Since phase detector 54, like phase detector 28, operates on the same set of samples (i.e., odd or even, but not both), loop filter 60 (like filter 38) is required to supply output samples at the symbol rate rather than at twice the symbol rate. This significantly reduces the complexity of the loop filters compared to what would be required for 2 sample per symbol implementations. 
     The output of multiplier 45, at the input to carrier recovery network 50, is a complex double sideband suppressed carrier AM signal centered at a frequency -2f c  -2Δ. The output of VCO 62 in the carrier loop is a signal approximately equal to 2Δ. To produce this signal at the output of VCO 62, a signal 1/4 f sr  is added to the carrier loop via multiplier 64 to cancel the 2f c  component. The complex signal 2Δ at the output of VCO 62 is translated to an output signal Δ of carrier recovery network 50 by means of a divide-by-2 frequency divider 70. Output signal Δ is a tone (without a frequency spectrum) representing the carrier phase/frequency offset. 
     The VSB output signal from ADC unit 16 is applied to an input of multiplier 74 after being delayed by a unit 72, which compensates for the signal delay through filters 20 and 22. The output of delay unit 72 is a complex near baseband symmetrical double sideband VSB signal as depicted by the frequency spectrum diagram adjacent to block 72. This signal is shifted to closer to baseband by multiplier 74, which is clocked at 1/8 f sr  to produce a near baseband upper VSB sideband at its output, as depicted by the frequency spectra diagram at the output of multiplier 74. The near baseband VSB signal from the output of multiplier 74 is applied to one input of a complex multiplier 71, and the output (offset representative) signal Δ from the output of carrier recovery network 50 is applied to another input of multiplier 71. The function of multiplier 71 is to substantially cancel the offset Δ in the VSB signal so that a baseband VSB signal results. 
     A complex demodulated VSB signal appearing at the output of multiplier 71 should be at baseband and often is. However, in practice this signal may contain residual phase offsets that may have to be compensated for. This is accomplished by an equalizer 75, which may be of a known configuration. Equalizer 75 compensates for channel disturbances as known and produces an equalized output signal that is decoded by unit 76 and processed by an output processor 78. Decoder 76 may include, for example, trellis decoder, de-interleaver, Reed-Solomon error correction, and audio/video decoder networks as known. Output processor 78 may include audio and video processors and audio and video reproduction devices. 
     Carrier recovery can also be accomplished using a single band edge of the input signal, as follows. Lower band edge filter 22 produces a double sideband AM output signal with a carrier frequency of -f c  +1/4 f st . This is accomplished by setting the input to multiplier 45 from filter 20 to a value of unity. This may be done by placing a multiplexer in the signal path between the output of filter 20 and the upper input of multiplier 45. One input of the multiplexer receives the output signal from filter 20, and another input receives a unity value signal. The latter signal is conveyed to the input of multiplier 45 in response to a control signal applied to a control input of the multiplexer. The output signal from upper band edge filter 20 is decoupled from multiplier 45 when the unity value signal is employed. In a system using two band filters, timing and carrier lock may occur at about the same time. In a system using only one band edge filter, carrier lock may occur after timing lock, depending on a variety of factors such as noise, loop gain and loop bandwidth. 
     If timing lock has been established by network 30, then f st  =1/4 f sr , and the f st  component of the carrier can be removed by shifting the AM output signal of filter 22 with a -1/8 f sr  spectral shifter. This spectral shifting is accomplished in FIG. 1 by changing the illustrated 1/4 f sr  clock input to multiplier 64 to a -1/8 f sr  clock. After such spectral shifting, the AM signal carrier at the output of multiplier 52 will be at a frequency -f c  (in contrast to -2f c  in the case of the double band edge method discussed previously). 
     In this embodiment the AM signal carrier frequency f c  can be easily driven to -1/4 f sr  by using a phase detector similar to unit 54 used for the double band edge carrier recovery example and forcing every other sample of the real component to zero. The output of the phase detector is integrated by a low pass loop filter which drives a VCO in a loop which acts to drive the AM signal carrier frequency to -1/4 f sr  in a manner analogous to that described above in connection with the double band edge carrier recovery example. 
     In the described embodiments, timing recovery (lock) may be achieved even in the presence of a carrier offset, and timing lock does not rely on sync components in the VSB signal for this purpose. Moreover, carrier recovery is accomplished without reliance on a pilot component. The choice of operating phase detectors 28 and 54 at the midpoint of the Nyquist region is one possible implementation. However, the phase detectors could also be operated at baseband with the same results. The use of the negative frequency spectrum is arbitrary. The positive spectrum could also have been used with analogous results, by using a different implementation of the described circuits. For example, conjugate filters 20 and 22 would be used and the input to multiplier 64 would be -1/4 f sr .