Patent Publication Number: US-7715815-B2

Title: Integrated tracking filters for direct conversion and low-IF single conversion broadband filters

Description:
RELATED APPLICATIONS 
   This application is a continuation of application Ser. No. 10/442,260, filed May 21, 2003, now U.S. Pat. No. 7,336,939, which is incorporated by reference herein in its entirety. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to radio frequency (RF) receivers, and more particularly, to RF receivers with tracking filter banks for single conversion broadband filters. 
   2. Description of the Related Art 
   Direct-conversion, or homodyne, receivers are popular for many communications applications because of their simplicity and low power. They do not require intermediate-frequency (IF) filters, which are often costly, and need only one frequency conversion stage and one local oscillator (LO). 
   Direct-conversion receivers rely entirely on quadrature mixing to obtain the necessary image rejection. Single-conversion low-IF receivers for broadband communication systems, such as direct broadcast satellite (DBS), broadcast or cable television (CATV), may rely on some combination of quadrature conversion and pre-selection with tracking or switched filters. 
   Double-conversion receivers for these applications rely more on IF filtering for image rejection. Because the IF frequencies can be fixed, very sharp filters, such as surface acoustic wave (SAW) filters, need to be used. However, these filters are relatively costly. 
   Difficulties in integrating tracking filters and in achieving the necessary degree of image rejection through quadrature balance have heretofore prevented the implementation of integrated direct-conversion receivers for broadcast television and CATV. Integrated single-conversion low-IF tuners generally do not include the tracking filters. 
   A typical conventional single-conversion tuner is illustrated in  FIG. 1 . As shown in  FIG. 1 , the conversion tuner includes a bank of bandpass filters  101   a ,  101   b ,  101   c  (in this example, three filters) that receive RF input. The output of the bandpass filters  101  is fed into a low-noise amplifier  104 , and then to a mixer  109 . The tuner also includes a phase-lock loop (PLL)  120 , which is comprised of a phase-lock loop controller  105 , and voltage controlled oscillators (VCO&#39;s)  106   a ,  106   b ,  106   c . Each VCO  106  is matched to a corresponding bandpass filter  101 . The output of a phase-lock loop controller  105  controls the filters  101  and the voltage control oscillators  106 . The outputs of the VCOs  106  are also inputted into the mixer  109 . 
   As further illustrated in  FIG. 1 , the output of the mixer  109  is received by a variable gain amplifier (VGA)  110 , which is placed in a feedback loop that includes a power detector  111 . The output of the variable gain amplifier  110  then is inputted to a downstream demodulator (not shown in  FIG. 1 ). 
   Thus, each tracking filter  101  is slaved to a corresponding voltage-controlled oscillator (VCO)  106   a ,  106   b ,  106   c , which produces the local oscillator (LO) for the first frequency conversion when that filter  101  is selected. The choice of filter  101  and VCO  106  depends on the frequency of the desired channel, and is determined by a digital command. By using matched components, such as inductors, capacitors, and voltage-variable capacitors (varactors) in the VCO  106  and filter  101 , the center frequency of each filter  101  can be matched to the frequency of the corresponding VCO  106 . This arrangement is illustrated in  FIG. 2 . 
   As shown in  FIG. 2 , each bandpass filter  101  can include an inductor L 205 , a capacitor C 207 , a varactor diode D 208 , and a capacitor C 206 , connected as shown. Choke inductors L 204  are used to prevent leakage of RF signals back upstream. As further shown in  FIG. 2 , the voltage control oscillator  106  includes active components  202 , and passive components L 210 , C 212 , D 213 , and C 211 , connected as shown. A choke L 209  similarly prevents leakage of RF signals back upstream from the VCO  106 . 
   A secondary problem in direct-conversion tuners are the responses at the LO harmonics, as shown in the bottom graph of  FIG. 1 . For a narrowband communication system, where the input spectrum spans less than one octave, this is not a problem. There are no signals to be received at the frequencies corresponding to the LO harmonics. Any unwanted signals can in principle be removed with a fixed filter. But for wideband communication systems such as CATV and terrestrial broadcast television, this issue must be addressed. For example, for a TV channel at 50 Mhz, there is an unwanted harmonic response at 150 Mhz, as shown in  FIG. 1 . 
   SUMMARY OF THE INVENTION 
   The present invention is directed to integrated tracking filters for direct conversion and low-IF single conversion broadband filters that substantially obviates one or more of the problems and disadvantages of the related art. 
   There is provided a radio frequency (RF) tuner including a programmable tracking filter bank receiving an RF input and outputting a filtered RF signal. A mixer stage receives the filtered RF signal and outputs a first quadrature component of the filtered RF signal and a second quadrature component of the filtered RF signal. Two variable gain amplifiers receive the first and second quadrature components and output amplitude-controlled I and Q components of the filtered RF signal. In one embodiment, the programmable tracking filter bank includes a plurality of tank circuits each connected to the RF input through an impedance. Each tank circuit includes an inductor and a capacitor connected in parallel, thereby forming an LC network, and a plurality of switched capacitors in parallel with the LC network and switched in and out of the tank circuit by programmable switches. In another embodiment, the programmable tracking filter bank includes a plurality of peaked low-pass circuits each connected to the RF input through an impedance. Each peaked low-pass circuit includes a capacitor connected to ground, and a plurality of switched capacitors in parallel with the capacitor and switched in and out of the peaked low-pass circuit by programmable switches. 
   In another aspect there is provided a radio frequency (RF) tuner including a programmable tracking filter bank receiving a differential RF input and outputting a filtered RF differential signal. A mixer stage receives the filtered differential RF signal and outputs a first quadrature component of the filtered differential RF signal and a second quadrature component of the filtered differential RF signal. Two variable gain amplifiers receive the first and second quadrature components, and output amplitude-controlled I and Q components of the filtered differential RF signal. 
   In another aspect there is provided a radio frequency (RF) receiver including a tuner receiving an RF input and outputting amplitude controlled I and Q components of the RF input. A demodulator converts the I and Q components to a TV channel signal. A power detector receives the demodulated TV channel signal, and outputs a control signal to a filter control circuit. The filter control circuit controls the tuner. The tuner includes a programmable tracking filter bank receiving the RF input and outputting a filtered RF signal based on output of the filter control circuit. A mixer stage receives the filtered RF signal and outputs a first quadrature component of the filtered RF signal and a second quadrature component of the filtered RF signal. Two variable gain amplifiers receive the first and second quadrature components and output the amplitude-controlled I and Q components of the filtered RF signal. 
   Additional features and advantages of the invention will be set forth in the description that follows. Yet further features and advantages will be apparent to a person skilled in the art based on the description set forth herein or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings. 
   It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS/FIGS. 
     The accompanying drawings, which are included to provide a further understanding of the exemplary embodiments of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings: 
       FIG. 1  shows a conventional single conversion tuner with a tracking filter bank. 
       FIG. 2  shows tracking filters matched to a voltage controlled oscillator. 
       FIG. 3  shows a direct conversion tuner of the present invention. 
       FIG. 4  shows one embodiment of a splitter and tunable filter bank. 
       FIG. 5  shows an example of one element of a differential filter bank. 
       FIG. 6  shows filter centering through baseband (or low-IF) power detection. 
       FIG. 7  shows another embodiment of a splitter and tunable filter bank using peaked low-pass sections. 
       FIG. 8  shows a frequency response of the peaked low pass sections of  FIG. 7 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Reference will now be made in detail to the embodiments of the present invention, examples of which are illustrated in the accompanying drawings. 
   This invention describes techniques for implementing tunable tracking filters for RF applications, suitable for conventional integrated circuits, with minimal off-chip components. In particular, this invention discloses a technique for implementing tunable tracking filters suitable for image and LO harmonic response rejection in integrated broadband tuners. 
   A direct-conversion tuner  350  of the present invention is shown in  FIG. 3 . In this case, the IF frequency is zero, so tracking filters cannot be used to provide image rejection. Quadrature mixers with adequate balance are therefore necessary. 
   As shown in  FIG. 3 , the tuner  350  according to the present invention includes a low-noise amplifier (LNA)  301  receiving RF input. Typical RF input is a television signal. In the U.S., the signal occupies 50-860 megahertz spectral range and carries a plurality of 6 MHz bandwidth channels. The output of the low-noise amplifier  301  is fed into a bank of programmable tracking filters  311 , then to two mixers  302   a ,  302   b . The purpose of the programmable filter bank  311  is to suppress RF at harmonics of the phase-lock loop  305 , so that the overall circuit has no response at those frequencies. A phase-lock loop (PLL)  305  outputs a waveform to the mixer  302   a  and to a 90° phase shifter  304 . The phase shifter  304  outputs a shifted waveform from the PLL  305  to the mixer  302   b . Collectively, the PLL  305 , the phase shifter  304  and the mixers  302   a ,  302   b  may be referred to as a mixer stage  325 . 
   The outputs of the mixers  302   a ,  302   b  are fed into low-pass filters  306   a ,  306   b  respectively. A typical low-pass filter  306  might have a bandwidth of 3 MHz, for the U.S. standard of 6 MHz per TV channel. The low-pass filters  306   a ,  306   b  output their filtered signals to variable gain amplifiers (VGAs)  308   a ,  308   b . The variable gain amplifiers  308   a ,  308   b  are controlled by power detection circuits  310   a ,  310   b  respectively, arrayed in feedback loops. The VGAs  308   a ,  308   b  also output the I and the Q quadrature components of the TV signal. 
     FIG. 4  illustrates the bank of tunable filters  311  of one embodiment of the present invention. As shown in  FIG. 4 , the bank of tunable filters  311  includes a plurality of impedances  401   a ,  401   b  . . .  401   n , which may be, e.g., inductors. Each tunable filter also includes a tank circuit (i.e., a band pass filter)  430 , which does not need to be as sharp a filter as a filter needed for image rejection. Note that each tank circuit  430  covers a particular range of the TV spectrum. Each tank circuit  430  includes an inductor L 406 , a capacitor C 402 , and switchable capacitors C 403 , C 404 , C 405  (only three are shown in  FIG. 4 , but more or fewer may be used), which are switched in and out of the circuit by switches (e.g., transistors) M 407 , M 408 , M 409 . Note that in a typical circuit, the inductor L 406  is usually an on-board inductor, whereas the rest of the tank circuit  430  can be manufactured as an integrated circuit. Generally, however, the inductors L 406  may be discrete, or printed. 
   The outputs V OUTA , V OUTB  . . . V OUTN  are outputted through switches S 1 , and are summed and outputted to the mixer stage  325  of  FIG. 3 . Note that the output V OUT0  represents the higher frequencies, and in a typical TV tuner circuit harmonics higher than second or third are usually not a concern. Thus, there is usually no need for filters above approximately 280-300 MHz. The frequency-selective circuit of each tank circuit  430  is a single parallel LC tuned circuit. The inductor L 406  may be on-chip or off-chip, as noted above, although off-chip inductors generally are available with higher Q factors and give better performance. The capacitance element C 402  is tuned on-chip. The small capacitive elements C 403 , C 404 , C 405  may be connected or disconnected from the tuned circuit using integrated transistor switches (M 407 , M 408 , M 409 ). There will be some amount of fixed parasitic capacitance, both on-chip and off-chip. There may also be some amount of intentional capacitance, either on-chip or off-chip. 
   The inputs of the tank circuits  430  are connected to a common input through impedances  401  (e.g., inductors). This prevents interaction between the tuned circuits. This arrangement is essentially a frequency multiplexor, with one common input V in  and multiple outputs V OUTA , V OUTB , . . . V OUTN , such that each output passes a different frequency band. Generally, the tank circuits  430  are tunable over overlapping ranges covering the entire input range. 
   If the tank circuits  430  are being used for image-rejection, they must provide sufficient attenuation at an offset from the desired frequency equal to twice the IF frequency, for all input frequencies. If they are being used for rejection of harmonic responses, they need only provide sufficient attenuation at the unwanted harmonic of the input frequency. For instance, for a TV channel at 62 Mhz, the harmonic of primary concern is at 186 Mhz, thus, it is most important to reject input at 186 Mhz, in this case. As noted above, no filtering is necessary for frequencies greater than the maximum input frequency divided by the order of the lowest significant harmonic response. This significantly simplifies the architecture of the programmable filter bank  311 . 
   In  FIG. 4 , the multiple outputs V OUTA -V OUTN  are recombined to a single signal driving the first mixing stage  302   a ,  302   b  with switches S 1   a , S 1   b  . . . S 1   n . Each tank circuit  430  output may be connected or disconnected from the first mixing stage  302   a ,  302   b  using, e.g., an integrated transistor switch. 
   Alternatively, each tank circuit  430  output can drive an independent amplifier or mixer stage, whose outputs will be combined to drive the subsequent stages of the tuner  350 . 
   The number of tank circuits  430  in the filter bank  311  is determined by the total frequency range that must be covered, and by the tuning range of each tank circuit  430 . The inductance L of the inductor L 406  and the maximum and minimum capacitance C max  and C min  of the capacitor C 402  that can be realized in parallel determine the tuning range of a given tank circuit  430 . C min  is determined by the parasitic capacitances on and off the chip, including printed circuit board (PCB) wiring traces, inductor parasitics, IC bond pad parasitics, and the parasitic capacitance of the integrated transistor switches (M 407 -M 409 ) that are used to vary the capacitance. 
   Let C 0  (not designated in  FIG. 4 ) be the fixed parasitic capacitance. Let C SW  (also not designated in  FIG. 4 ) be the parasitic capacitance of a unit-sized transistor switch. The size of each switch must be large enough so that the Q of the tuned circuit, and consequently the frequency response of the corresponding tank circuit  430 , are not unacceptably degraded by the switch resistance. Let W be the minimum acceptable switch size relative to a unit switch. Then its parasitic capacitance is WC SW . W is in turn approximately proportional to the switched capacitance. Therefore, 
   
     
       
         
           
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   Thus, it is desirable that k, the proportionality factor between the switched capacitance and the parasitic capacitance of a transistor switch be large enough to connect or disconnect it from the circuit without excessive Q degradation. This requirement favors high-speed integrated circuit processes with very small parasitic capacitances for a given amount of switch conductance. Note that either bipolar or field-effect transistors may be used as switches. The long recovery time from saturated operation if bipolar transistors are used is not relevant, since the rate at which the tank circuit  430  center frequency will need to be tuned is typically quite slow. 
   Thus, the tracking filters  311  described herein may be used to eliminate image responses as well as responses at the harmonics of the LO frequency, which cause undesirable interference in the RF receiver. For example, in the case of a channel at 50 Mhz, the tracking filters  311  eliminate the responses at the harmonics of 50 Mhz, particularly the odd harmonics (e.g., 150 Mhz, etc.) 
     FIG. 5  illustrates an element of the filter bank  311  in differential form. Although the differential version requires additional components, it may be preferred for superior distortion performance and crosstalk immunity. As shown in  FIG. 5 , the differential form of each of the tracking filters  530  that can be used and the filter bank  311  includes impedances  501  and  512 , which receive the input signals Vin+ and Vin−. The signal then is inputted into the tracking filter  530  that includes an inductor L 506 , a capacitor C 502 , and a plurality of capacitors C 503 , C 504 ; C 505 , C 506 ; and C 507 , C 508 , which are switched in and out of the circuit to control the center frequency of the circuit by transistors M 509 , M 510 , M 511 , respectively. The outputs of the filter  530  are the V OUTA + and V OUTA − as shown in  FIG. 5 . 
   Further with reference to  FIG. 4 , an additional requirement is that the center frequency of each tank circuit  430  be essentially the same as the desired channel. Because of component tolerances, it may not be possible to know in advance how many of the switched capacitance elements C 403 , C 404 , C 405  must be connected for a given center frequency. In this case, some form of feedback control is necessary to tune the tank circuit  430  properly. This may be accomplished using the arrangement illustrated in  FIG. 6 , where the baseband power detector, generally present in most communication receivers as part of the automatic gain control function, indicates the amplitude of the desired channel downstream of all tuner processing. The tunable filter bank  311  can be swept over a sufficient range, and the sweep is concluded when the baseband power detector  604  indicates a maximum signal amplitude, corresponding to a well-centered filter. 
     FIG. 6  illustrates how the tuner  350  of the present invention is used in an overall receiver. As shown in  FIG. 6 , the RF input is received by the tuner  350 . The tuner  350  then outputs the I and the Q components of the signal to a baseband or low-IF processing circuit  602  (i.e., a demodulator). The demodulated signal, which represents a single TV channel, is then fed further downstream. A power detect circuit  604  is used in a closed loop manner to center the filter bank  311  of the tuner  350 . The power detect circuit  604  detects the power at the desired TV channel. A filter control circuit  605  controls the gates of the transistors of the tracking filters, so as to maximize the power of the desired channel detected by the power detect circuit  604 , as shown in  FIGS. 4 ,  5  and  7 . 
     FIG. 7  illustrates an alternative circuit configuration of the filter bank  311  that uses peaked second-order low-pass sections  730 , which can provide the desired filtering action while saving one (in the case of a single-ended section) or two (in the case of a differential section) inductors per section. As shown in  FIG. 7 , the RF input is received through a resistor R 701 . Each section  730  includes a capacitor C 710 , and capacitors C 702 , C 703 , C 705  (three in this example), which are switched in and out of the circuit by programmable switches (e.g., transistors) M 707 , M 708  and M 709 . The filter bank  311  shown in  FIG. 7  functions similarly to the filter bank of  FIG. 4 , except that it does not use an extra inductor and may be made more compact and better integrated into a single IC.  FIG. 8  shows a frequency response of the peaked low pass sections of  FIG. 7 . In  FIG. 8  shows the peaked low pass responses at a certain frequency (e.g., f 0 =50 Mhz) of three sections. One response would be selected as an input to the mixer. Alternatively, with multiple mixers, each response goes to a particular mixer. 
   In the case of the peaked low pass architecture, it is preferable that the impedance of integrated circuit at each tank circuit  430  output be capacitive, to facilitate a high Q frequency response with low loss and good selectivity. 
   Thus, as discussed above, the present invention permits centering using a baseband power detector. Furthermore, the present invention permits the use of a filter bank with switched tuning implemented on-chip. Additionally, the present invention permits filter outputs to be connected to downstream signal path with switches, or multiple filter outputs driving multiple amplifier or mixer stages, which are then combined. Thus, the present invention permits reducing the number of external components, does not require discrete varactors, and does not rely on critical component matching. 
   CONCLUSION 
   While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail can be made therein without departing from the spirit and scope of the invention. 
   The present invention has been described above with the aid of functional building blocks and method steps illustrating the performance of specified functions and relationships thereof. The boundaries of these functional building blocks and method steps have been arbitrarily defined herein for the convenience of the description. Alternate boundaries can be defined so long as the specified functions and relationships thereof are appropriately performed. Also, the order of method steps may be rearranged. Any such alternate boundaries are thus within the scope and spirit of the claimed invention. One skilled in the art will recognize that these functional building blocks can be implemented by discrete components, application specific integrated circuits, processors executing appropriate software and the like or any combination thereof. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.