Patent Publication Number: US-9413414-B2

Title: High isolation signal routing assembly for full duplex communication

Description:
RELATED APPLICATIONS 
     This application is continuation of application Ser. No. 13/375,161, filed Dec. 13, 2011, which is a continuation-in-part of application Ser. No. 12/459,242, filed Jun. 29, 2009, now issued as U.S. Pat. No. 8,077,639, which is a continuation-in-part of international application serial number PCT/US2007/026459, filed Dec. 29, 2007, now abandoned, which claims priority to U.S. provisional application Ser. No. 60/877,995, filed Dec. 29, 2006, expired, and each of said applications is herein incorporated by reference and priority is claimed to each of said applications. 
    
    
     TECHNICAL FIELD 
     The present invention relates to wireless transceivers that operate in full duplex mode providing the simultaneous transmission and reception of radio signals. In particular, but not exclusively, the present invention relates to wireless transceivers that are provided with a means to isolate signals transmitted by the transmitter of the wireless transceiver and received by a receiver of the wireless transceiver. 
     BACKGROUND OF THE INVENTION 
     Modern wireless communication, radar and radio frequency identification (RFID) systems often operate under full duplex operation. A wireless transceiver comprises of a local transmitter and a local receiver. Full duplex operation occurs when a local transmitter is actively transmitting RF signals during the same time that a local receiver is detecting RF signals and/or backscatter from the surrounding environment. The local transmitter and local receiver are typically in close proximity to one another and are often placed within a common enclosure. It is also desired to operate the full duplex system using a monostatic configuration, namely a configuration that uses a single antenna common to both the local transmitter and local receiver. In a typical transceiver, the transmitted and received signals are typically routed to and routed from the single antenna using a duplexing filter, circulator or directional coupler. 
     It is known that the operation of a local receiver during the time that a local transmitter is transmitting creates receiver problems as the transmitter energy leaks, couples and/or reflects into the receiver resulting in corruption, distortion, saturation and/or desensitization within the receiver. In some cases, a duplexing filter may be used to isolate the transmitted energy from the receiver if the transmitter and receiver are configured to operate at two different RF carrier frequencies that allow the duplexing filter to provide the required isolation between the transmitter and receiver. If the system is designed to operate with the transmitter and receiver using the same RF carrier frequency or with different transmit and receive frequencies that are close in RF carrier frequency such that the duplexing filter can not adequately provide the required isolation, then a circulator or directional coupler is typically used to isolate the transmitted signal from entering the receiver. Depending on the isolation performance of the circulator or coupler, the system performance may degrade when a portion of the transmitted energy leaks into the receiver. 
     Circulators and directional couplers are three and four port devices that are used to route RF and microwave signals between various ports within the component. A RF or microwave signal entering the circulator or coupler is expected to exit at a desired port(s) where one port is isolated from the incident signal. In practice, a portion of the incident signal leaks or couples to the isolated port. The ratio of the undesired leakage to the incident signal is often referred to as the isolation of the device. When referring to directional couplers, the isolation term is sometimes referred as the coupler directivity which is defined as the (dB) difference between the isolation and the coupling value of the directional coupler. 
     A basic circulator,  1 , is a three-port device that provides primary signal transmission between pairs of ports. A symbolic diagram of a circulator  1  is shown in  FIG. 1 . Signals are routed between pairs of ports in the direction of circulation arrow  5 . For this example, the circulation arrow  5  in  FIG. 1  shows a clockwise direction for signal paths. This circulation arrow  5  is used in the technical literature as a symbolic reference to the direction of signal paths within the circulator. Circulators can be manufactured to have either clockwise or counter-clockwise signal directions. A signal  6  entering the input port  2  will exit through the desired output port  3  following the clockwise circulation arrow  5 . Ideally, the signal  7  leaving the circulator  1  will have the same magnitude level as the input signal  6 . In practice, the signal  7  leaving the circulator  1  will have some reduction in amplitude due to losses and mismatches that occur inside the circulator. The signal  7  leaving the circulator  1  will also have a phase shift relative to the input signal  6 . Ideally, no portion of the input signal  6  should leave the third port  4 . This third port  4  is the isolated port. In practice, the isolated port  4  will have a signal level  8  reduced by approximately 20 dB when measured relative to the input signal  6 . In practice, junction circulators typically have a minimum isolation of 20-25 dB and lumped element circulators typically have a minimum isolation of 13 dB. 
     The circulator  1  can also be configured to operate with the input signal entering port  3  and exiting through port  4 . In this case port  2  is the isolated port. The circulator  1  can also be configured to operate with the input signal entering port  4  and exiting through port  2 . In this case port  3  is the isolated port. 
     A typically transceiver application using a monostatic antenna configuration is shown in  FIG. 2 . In the case a local transmitter  9  generates a transmitted signal  6  that enters the input port  2  of the circulator  1 . The circulator  1  routes the signal to the common port  3 . The signal  7  leaves the circulator at the common port  3  and enters the antenna  12 . Any received signal  10  captured by the antenna  12  from the surrounding environment enters the common port  3  of the circulator  1  and is routed to the output port  4 . The desired received signal  13  leaving the output port  4  enters the local receiver  11 . If the system uses a local transmitter  9  and local receiver  11  that are operating simultaneously, then a portion of the signal  6  from the active transmitter  9  may couple or leak through the circulator  1  and enter the active receiver  11 . This undesired coupled signal  8  may reduce the performance of the receiver  11 . 
     For certain applications where additional losses in the receive path will not effect the required system performance, it is possible to replace the circulator with a directional coupler. In this configuration, the directional coupler is used to route the signals between the transmitter  9  to the antenna  12 , and from the antenna  12  to the receiver  11 .  FIG. 3  shows a monostatic antenna configuration using a directional coupler  21 . In this configuration, the directional coupler  21  is positioned in order to transfer the signal  6  emitted from the local transmitter  9  to the antenna  12 . The directional coupler is typically a four-port device where one of the ports are terminated using a resistive termination  25 . The termination  25  is often matched to the characteristic impedance of the system, which is typically 50 ohms. In some cases, the termination  25  is included internally to the directional coupler which effectively makes the device into a three port component. A three port description for the directional coupler will be used throughout this disclosure. In the configuration shown in  FIG. 3 , the termination  25  is used to absorb energy coupled from the incident signal  6 . Ideally no portion of the incident signal  6  should leave port  24  which connects the directional coupler  21  to the receiver  11 . For signal reception from the surrounding environment, any received signals  10  captured by the antenna  12 , will enter the directional coupler  21  at common port  23 . A portion of the received signal  10  will be coupled to the output port  24 . The desired coupled signal  13  will then enter the receiver  11 . As the desired signal  13  is coupled to the output port  24  the amplitude level of the desired signal  13 , will be reduced by the coupling factor. In some full duplex systems, such as passive UHF RFID systems, this additional loss in received energy does not create difficulties when recovering the received information as these systems are generally forward link limited. Problems may occur when a portion of the incident transmitter signal couples or leaks into the receiver through the directional coupler which in turn may reduce receiver performance. In this case, a portion of the transmitted signal,  6 , may couple or leak through the directional coupler  21  and exit the output port  24 . This undesired coupled signal  8  will enter the receiver  11  and may reduce the performance of the receiver  11 . 
     In both configurations, shown in  FIG. 2  and  FIG. 3 , it is important that the receiver not be desensitized by any signal(s) coming from the system. Signals that could desensitize the receiver include signals received by the antenna and signals that leak or couple over from the transmit channel. If the signal received by the antenna from the surrounding environment is the signal of interest, then it is assumed that the system has been designed as not to desensitize the receiver when this signal is present. Therefore undesired receiver desensitization may occur when signals leak from the local transmitter into the local receiver. It is well known in industry, that fabricating a circulator with very high isolation (&gt;30 dB) is often difficult and expensive. A typical junction circulator may have 20 dB isolation resulting in 1% of the transmit energy leaking into the receiver channel. This leakage signal may greatly affect the performance of the receiver in a full duplex system. As an example, a typical RFID system that uses a transmitter with an output power of 1 watt (+30 dBm) and a circulator with an isolation of 20 dB would have an unwanted signal entering the receiver of 10 mwatt (+10 dBm). This level of undesired signal would typically saturate and/or desensitize a low power mixer placed in the front end of the receiver. 
     DISCLOSURE OF INVENTION 
     It is an object of the present invention to provide a two way duplex wireless communication signal routing assembly wherein the channel to the channel isolation is improved over prior art arrangements. In particular, the present invention relates to a signal routing assembly, a full duplex transceiver routing assembly and a full duplex transceiver routing assembly including carrier modulation. The signal routing assembly provides high isolation between two separate transmission signal paths utilizing a common port and typically configured to provide high isolation in the direction from the transmit channel transmission path to the receive channel transmission path in a full duplex system. The signal routing assembly allows the two transmission paths to operate using the same carrier frequencies. The signal routing assembly also allows the two transmission paths to operate using different transmit and receive frequencies. In an advantageous application the two different frequencies are close in frequency and are therefore inadequately filtered using a duplexing filter. 
     Briefly stated, the present invention provides a wireless communication device for effecting two way full duplex wireless communications, where the signal routing assembly accepts the first transmission signal at the input and outputs a substantial portion of this signal at the common port of the signal routing assembly and a second transmission signal received at the common port from the surrounding environment or other parts of the system is routed through the signal routing assembly and a portion of the second transmission signal is delivered to output of the signal routing assembly. Relative to the output of the signal routing assembly, leakage signals from the first transmission signal are terminated inside the signal routing assembly. In a typical system, a transmitter produces the first transmission signal and the signal routing assembly delivers a substantial portion of this signal to the common port of the signal routing assembly that is typically connected to the antenna. In full duplex operation, second transmission signals received at the antenna from the surrounding environment are routed to the receiver through the signal routing assembly from the common port to the output where a at least a portion of a second transmission signal is delivered to the receiver. The signal routing assembly cancels a substantial portion of the transmitter leakage signal from entering the receiver. 
     In an embodiment of the present invention, the signal routing assembly includes a signal divider receiving the first transmission signal and dividing the first transmission signal into first and second divided transmission signals having substantially equal amplitudes and a first relative phase shift therebetween. First and second routing devices are provided each having at least first, second and third ports, and being configured to simultaneously deliver a signal at the first port to the second port and another signal at the second port to the third port each at functionally operative levels. The first and second routing devices receive the first and second divided transmission signals at the first ports and routes them to the second ports producing a first and second divided transmission output signal respectively. The first and second routing devices simultaneously output a first and second transmission leakage signal at the third ports respectively. The first and second routing devices receive, when present, third and fourth divided transmission signals at the second ports and routes them to the third ports producing third and fourth divided transmission output signals respectively. Further provided is a signal divider/combiner having first and second divider/combiner ports configured to combine the first and second divided transmission signals with a second relative phase shift therebetween to output a substantial portion of the first transmission signal to the common port of the signal routing device. The signal divider/combiner receives, when present, a second transmission signal and dividing the second transmission signal into third and fourth divided transmission signals having substantially equal amplitudes and a second relative phase shift therebetween. Further provided is a signal combiner having first and second combiner inputs receiving first and second transmission leakage signals and, when present, third and fourth divided transmission output signals. The signal combiner is configured to introduce a third relative phase shift into at least one of the signals applied to the combiner inputs such that; the first and second transmission leakage signals have approximately 180 degrees relative phase shift and arrive at approximately the same amplitude levels at the output to substantially cancel each other, and the third and fourth divided transmission output signals have approximately 0 degrees relative phase shift and arrive at approximately the same amplitude levels at the output to substantially combine with each other. 
     In an embodiment of the present invention the signal divider is optionally a quadrature hybrid. Alternatively, the signal divider may be embodied as an equal phase power dividing device with a phase shift introduced into one branch. 
     It is a feature of the present invention the signal combiner is optionally a quadrature hybrid. Alternatively, the signal combiner may be embodied as an equal phase power dividing device with a phase shift introduced into one branch. Such an equal phase power combiner will preferably include a resistive element in which undesired signals are dissipated. 
     It is a further feature of the present invention that the signal divider/combiner is embodied as a quadrature hybrid. Alternatively, the signal divider/combiner maybe embodied as an equal phase power divider/combiner with a phase shift introduced into one branch. 
     Yet another feature of the present invention is the use of circulators as the first and second routing devices. It is preferable that the first and second routing devices are electrically matched. Alternatively, one may embody the first and second routing devices as directional couplers. 
     It will be appreciated that any combination of the above noted embodiments of the signal divider, the signal divider/combiner, the signal combiner, and the routing devices may be used. Since two different examples of embodiments are discussed for each of the four devices, one will observe this yields sixteen combinations, the explicit recitation of which is unnecessary as such combinations will be understood. 
     It is a further feature of the present invention that in the signal routing assembly will deliver a substantial portion of the first transmission signal to the signal routing assembly common port where a substantial portion is in the range of 0.3 dB to 2.5 dB less in amplitude level relative to the first transmission signal when circulators are implemented as signal routing devices. Alternately, one may employ directional couplers as signal routing devices; therefore, a substantial portion is in the range of 0.2 dB to 4.0 dB. 
     It is a still further feature of the present invention that in the signal routing assembly will deliver a portion of the second transmission signal to the signal routing assembly output where a substantial portion is in the range of 0.3 dB to 2.5 dB less in amplitude level relative to the second transmission signal when circulators are implemented as signal routing devices. Alternately, one may employ directional couplers as signal routing devices; therefore, a substantial portion is in the range of 6.0 dB to 40.0 dB. 
     It is a still further feature of the present invention that the first and second transmission leakage signals produced by the first transmission signal substantially cancel each other such that a signal appearing at the output of the signal routing assembly is at least 22 dB below the amplitude level of the first transmission signal entering the signal routing assembly when circulators are used as signal routing devices. Preferably, this value will be at least 27 dB. Still more preferably, this value will be at least 37 dB. 
     Yet another feature of the present invention that the first and second divided transmission signals leaving the signal divider output ports may be amplified to increase the amplitude level of the divided transmission signals. 
     Yet still another feature of the present invention that the first and second divided transmission signals are modulated as to adjust the signal amplitude and/or phase. 
     Further features of the present invention include divider/combiner leakage cancellation configurations which compensate for leakage in the divider/combiner arising from the configuration of the signal divider/combiner producing a third transmission leakage signal, at the first divider/combiner port, which is a portion of the second divided transmission output signal and has an amplitude equal to an amplitude of the second divided transmission output signal multiplied by H and a phase shift −φH relative to the second divided transmission output signal, and the signal divider/combiner further producing a fourth transmission leakage signal, at the second divider/combiner port, which is a portion of the first divided transmission output signal and has an amplitude equal to an amplitude of the first divided transmission output signal multiplied by H and a phase shift −φH relative to the first divided transmission output signal. 
     A first embodiment of a divider/combiner leakage cancellation configuration includes a reflector device applied to the common port and configured to have a reflection coefficient R to reflect into the common port a portion of the substantial portion of the first transmission signal as a reflected signal of amplitude equal to an amplitude of the substantial portion of the first transmission signal multiplied by R and relative phase shift −φR. The configuration of the signal divider/combiner is so arranged as to divide the reflected signal into the first and second reflected divided signals having substantially equal amplitudes and a second relative phase shift therebetween, with the first and second reflected divided signals being respectively output at the first and second divider/combiner ports. The first routing device receives the first reflected divided signal and the third transmission leakage signal at the second port and produces, simultaneously at the third port a first reflected divided output signal and a third transmission leakage output signal. The second routing device receives the second reflected divided signal and the fourth transmission leakage signal at the second port and produces, simultaneously at the third port a second reflected divided output signal and a fourth transmission leakage output signal. The signal combiner has the first and second combiner inputs respectively receiving the third and forth transmission leakage output signals, and respectively receiving the first and second reflected divided output signals. The signal combiner is so configured as to combine the first and second reflected divided output signals at the transmission signal output with the third and fourth transmission leakage output signals to effect substantial cancellation of the third and fourth transmission leakage output signals. 
     The divider/combiner leakage cancellation configuration has the configuration of the reflector device set to have the reflection coefficient R and the relative phase −φR so as to effect the substantial cancellation of the third and fourth transmission leakage output signals by having R substantially equal H and −φR substantially equal to (−φH−90+2(φ8)) wherein:
         φ8 is a net electrical length of a portion of the common connecting line between said reflecting device and said common port of said signal divider/combiner.       

     A second embodiment of a divider/combiner leakage cancellation configuration includes a reflector device applied in a connection line between the second port of the first routing device and the first divider/combiner port of the signal divider/combiner and configured to have a reflection coefficient X to reflect into the second port of the first routing device a portion of the first divided transmission output signal as a reflected signal of amplitude equal to an amplitude of the first divided transmission output signal multiplied by X and relative phase shift −φX. The first routing device receives the reflected signal and the third transmission leakage signal at the second port and produces, simultaneously at the third port a reflected output signal and a third transmission leakage output signal. The second routing device receives the fourth transmission leakage signal at the second port and produces, simultaneously at the third port a fourth transmission leakage output signal. Finally, the signal combiner has the first and second combiner inputs respectively receiving the third and forth transmission leakage output signals, and the first combiner input receiving the reflected output signal. The signal combiner is so configured as to combine a portion of the reflected output signal at the transmission signal output with the third and fourth transmission leakage output signals to effect substantial cancellation of the third and fourth transmission leakage output signals wherein the configuration of the reflector device is set to have a reflection coefficient equal to X and the relative phase −φX so as to effect the substantial cancellation of the third and fourth transmission leakage output signals. 
     The second embodiment of a divider/combiner leakage cancellation configuration includes X being set substantially equal to 2H and −φX being set substantially equal to (−φH−90−2(φ4)+2(φ6)) wherein:
         φ4 is a net electrical length of a first connecting line connecting the second port of the first routing device to the first divider/combiner port;   φ4 is a net electrical length of a second connecting line connecting the second port of the second routing device to the second divider/combiner port; and   φ6 is a net electrical length of a portion of the first connecting line between the reflecting device and the second port of the first routing device.       

     A third embodiment of a divider/combiner leakage cancellation configuration is constructed and functions as does the second embodiment with the exception that the reflector device is applied in a connection line between the second port of the second routing device and the second divider/combiner port of the signal divider/combiner. 
     A fourth embodiment of a divider/combiner leakage cancellation configuration is constructed as a combination of the second and third embodiment and has a first reflector device applied in a connection line between the second port of the first routing device and the first divider/combiner port of the signal divider/combiner, and a second reflector device applied in a connection line between the second port of the second routing device and the second divider/combiner port of the signal divider/combiner so as to effect an imbalance resulting in cancellation of the third and fourth transmission leakage output signals. 
     The reflector devices in the above cancellation configurations are an open stub, a shorted stub, or a reactive component selected from the group consisting of a capacitor and an inductor. 
     A fifth embodiment of a divider/combiner leakage cancellation configuration includes a magnetic biasing device applied to one of the circulators so as to effect an imbalance in the first and second transmission leakage signals and third and fourth transmission leakage output signals resulting in cancellation of the first and second transmission leakage signals and third and fourth transmission leakage output signals at the received signal output. The magnetic biasing device is a magnetic device, a metallic device or a coiled wire carrying electrical current that is placed in the vicinity of the circulator. 
     The above, and other objects, features and advantages of the present invention will become apparent from the following description read in conjunction with the accompanying drawings, in which like reference numerals designate the same elements. The present invention is considered to include all functional combinations of the above described features and is not limited to the particular structural embodiments shown in the figures as examples. The scope and spirit of the present invention is considered to include modifications as may be made by those skilled in the art having the benefit of the present disclosure which substitute, for elements presented in the claims, devices or structures upon which the claim language reads or which are equivalent thereto, and which produce substantially the same results associated with those corresponding examples identified in this disclosure for purposes of the operation of this invention. Additionally, the scope and spirit of the present invention is intended to be defined by the scope of the claim language itself and equivalents thereto without incorporation of structural or functional limitations discussed in the specification which are not referred to in the claim language itself. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a prior art diagram of a circulator showing signal paths for desired and undesired signals that pass through the device; 
         FIG. 2  is a prior art diagram of a complete transceiver system using a circulator to route signals from the transmitter to the antenna and from the antenna to the receiver. Also shown is the undesired signal entering the receiver; 
         FIG. 3  is a prior art diagram of a complete transceiver system using a directional coupler to route signals from the transmitter to the antenna and from the antenna to the receiver. Also shown is the undesired signal entering the receiver; 
         FIG. 4  is a diagram of an embodiment of the routing device; 
         FIG. 5  is a diagram of an embodiment showing details of the routing device; 
         FIG. 6  is a diagram of an embodiment showing signal paths proceeding from the transmitter to the circulators; 
         FIG. 7  is a diagram of an embodiment showing signal paths proceeding from the circulators to the receiver and termination; 
         FIG. 8  is a graph of measured results for the isolation between the transmit channel to the receive channel; 
         FIG. 9  is a graph measured results for the isolation between the receiver channel to the transmit channel; 
         FIG. 10  is a diagram of an embodiment of the routing device using directional couplers as the routing device; 
         FIG. 11  is a diagram of an embodiment of the routing device using equal-phase power dividers and equal-phase power combiners that include a phase shift network; 
         FIG. 12  is a diagram of an embodiment of the routing device including modulators and amplifiers to modulate and amplify the input transmission signal; 
         FIG. 13  is a diagram of an embodiment showing signal paths proceeding from a leakage source to the receive channel; 
         FIG. 14  is a diagram of an embodiment showing the signal paths from a reflective device to the receive channel; 
         FIG. 15  is a diagram of an embodiment showing the signal paths from a reflective device to the receive channel; 
         FIG. 16A  is a diagram of an embodiment showing the reflective device configured as an open stub; 
         FIG. 16B  is a diagram of an embodiment showing the reflective device configured as a shorted stub; 
         FIG. 16C  is an embodiment showing the reflective device configured as a reactive lumped element; 
         FIG. 17  is a diagram of an embodiment of the routing device including magnetic bias device placed in the vicinity of the circulator; 
         FIG. 18  is a diagram of an embodiment of the routing device using power dividers as the routing device; and 
         FIG. 19  is a diagram of an embodiment showing signal paths from a common port to the two divided output ports. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Improvements in transmit to receive isolation for three-port signal routing devices can be provided using a combination of signal dividers with the proper signal phasing and conventional three port components that when properly connected will combine signals at the desired ports and cancel signals at the isolated ports. 
     The generalized construction of the three port signal routing device is shown in  FIG. 4 . The routing device  50  has one input port  51 , one common port  52  and one output port  53 . The input signal is received at the input port  51  and routed to the input  59  of a signal divider  54 . Input port  51  is typically connected to the local transmitter. Signal divider  54  divides the transmission signal into first and second divided transmission signals output at ports  60  and  61  and having substantially equal amplitudes and a first relative phase shift therebetween. The signal divider  54  is any of a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one output, or other device so functioning to divide a signal. 
     The first and second divided signals are routed to first and second routing devices,  55  and  56 , each having at least first, second and third ports. The divided signals enter the first ports and are routed to the second ports, the outputs of which are applied to the signal divider/combiner  57 . The signal divider/combiner  57  is any of a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one output, or other device so functioning to combine the signals from the second ports of routing devices  55  and  56 . The combined signal exits the divider/combiner  57  at port  66  and is routed to the common port  52 . Common port  52  is typically connected to an antenna for signal transmission and reception. Any signal entering the routing device  50  at the common port  52  is routed to port  66  of the signal divider/combiner  57  which divides the received signal into third and fourth divided transmission signals output at ports  64  and  65  and having substantially equal amplitudes and a first relative phase shift therebetween. 
     The third and fourth divided signals are routed to first and second routing devices,  55  and  56 . The divided signals enter the second ports and are routed to the third ports, the outputs of which are applied to the signal combiner  58 . The signal combiner  58  is any of a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one output, or other device so functioning to combine the signals from the third ports of routing devices  55  and  56 . The combined signal exits the combiner  58  at port  73  and is routed to the output port  53 . Output port  53  is typically connected to the local receiver. 
     The routing devices,  55  and  56 , are preferably matched circulators which provide some degree of isolation between the first ports and the third ports. Alternatively, the routing devices,  55  and  56 , are directional couplers. 
     The first and second routing devices,  55  and  56 , are devices intended to transfer a first signal from the first port to the second while simultaneously transferring another second signal entering the second port to the third while preventing the first signal from appearing at the third port. This is the idealized concept of such a routing device. However, in actual embodiments some of the first signal undesirably leaks through to the third port. The amount of this leakage is characterized by the isolation of the device wherein the greater the isolation (measured generally in decibels or dBs) is the higher the isolation value is. For the purposes of this disclosure the routing devices are characterized by transmission coefficients including:
         s 21  being a transmission coefficient from the first port to the second port;   s 32  being a transmission coefficient from the second port to the third; and   s 31  being a transmission coefficient from the first port to the third port;
 
wherein s 21  is greater than s 31 , and s 32  is greater than s 31 .
       

     For the purposes of this disclosure intended signal transfers are considered transfers at functionally operative levels meaning a level at which the signals transferred effect a desired function in the application of the device. Hence, applying this terminology to a simple switch transferring a signal, when the switch is on it would transfer a signal from an input to an output at a functionally operative level. If the switch is off, some leakage may occur resulting in a portion of the signal appearing at the output, this portion of the signal would not be considered to be at a functionally operative level since it would be attenuated to a level not intended to effect operation and not effecting a desired operation. 
     The signal combiner  58  has first and second combiner inputs and a received signal output connected to the receiver. The first and second combiner inputs are respectively connected to the third ports of the first and second routing devices,  55  and  56 , to accept the received signals from the common port  52 . The signal combiner  58  introduces a phase shift into signals applied to at least one of the first and second combiner inputs such that the received signals from the common port  52  are combined substantially in phase to produce the received signal at a received signal output which connects to the receiver. Transmission leakage signals which leak from the first ports to the third ports of the routing devices,  55  and  56 , are substantially phase shifted relative one another 180 degrees at the received signal output to substantially cancel each other. The signal combiner  58  may be a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter/combiner, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one of two inputs. 
     In the routing device  50 , connecting lines  62 ,  63 ,  67 ,  68 ,  69  and  70  interconnect the components and are described in more detail below. It is understood that components may be directly connected to each other and connecting lines omitted where feasible. In the preferred embodiment connecting lines  62  and  63  are electrically matched, connecting lines  67  and  68  are electrically matched and connecting lines  69  and  70  are electrically matched. However, it will be understood that it is not necessary that each of these pairs of lines be matched provided that overall phase shifts of and attenuations of signals are such that the transmitted signals are properly combined for transmission to the antenna and received signals are properly combined for transmission to the receiver. In order to provide adequate transmit channel to receive channel isolation, the overall phase shifts and insertion losses of the connecting lines or equivalents should present an overall phase shift and insertion loss introduced by connecting lines  62 ,  63 ,  69  and  70 , or their equivalents, present the transmission leakage signals of substantially equal amplitude and phase shifted relative one another about 180 degrees at the received signal output to substantially cancel each other. 
     In the preferred embodiment discussed below, improved isolation of the routing device  50  is achieved by the effective cancellation of the transmission leakage signal at the received signal output. The phase shifting of these undesired signals to effect cancellation should be such that transmit to receive isolation of at least 30 dB is achieved over a frequency range associated with the system use. More preferably, the insertion losses and phase shifts should effect matching resulting in at least 35 dB isolation over the frequency range. Still more preferably, the insertion losses and phase shifts should effect matching resulting in at least 40 dB isolation over the frequency range. Matching tolerances and effectiveness are discussed below. 
     It will be additionally appreciated from this disclosure that the phase shifts discussed herein are relative between the respective signals discussed and do not include multiples of 360 degrees electrical length difference that may exist in one connection over another. In other words and as merely an example, for the purposes of this disclosure, unless noted otherwise, a phase shift of 360 degrees or multiples thereof between signals is not considered to be a portion of a relative phase shift. Hence, a signal which is shifted 450 degrees relative another signal, is considered to be shifted 90 degrees for the purposes of this disclosure. Accordingly, it is understood that relative shifts and limitations related thereto recited herein do not exclude the addition of integer multiples of 360 degrees unless specifically stated. While it is preferable that electrical length differences of greater than 360 degrees are not introduced, such difference are not considered to be outside the scope of the present invention. 
     It will also be appreciated in view of this disclosure that practical production tolerances will result in slight differences in electrical characteristics between the connecting lines and between the first and second routing devices. Tuning elements and/or phase adjustment may be inserted along any connecting line in order to adjust the amplitude and phase of the signal traveling along the line. Tuning the signal may improve the isolation between the transmit channel and receive channel by compensating for any differences between the signal paths and components. Such tuning elements may include stubs or lumped components or other devices as are known by those skilled in the art. Additionally, for the purposes of this disclosure and unless stated otherwise, the connecting lines shown interconnecting components are not intended to exclude insertion of other components in those connecting lines for tuning, amplification or other purposes provided that the cancellation of the transmission leakage signals are achieved at the signal combiner  58 . 
     Referring to  FIG. 5 , details of a preferred embodiment of the present invention are described herein wherein the generalized internal components of the routing device  50  as disclosed above are embodied in devices used in implementation of the preferred embodiment. It is understood that the above discussion with relation to the generalized components and interconnections shown in  FIG. 4  applies to the preferred embodiment shown in  FIG. 5 . 
     In  FIG. 5  the routing device  50  uses three quadrature hybrids, an input quadrature hybrid  110 , a common quadrature hybrid  111  and an output quadrature hybrid  112 , and first and second circulators,  100  and  101 , connected in such a way as to prevent unwanted transmission energy from the transmitter from entering the receiver. The input quadrature hybrid  110 , common quadrature hybrid  111  and output quadrature hybrid  112  need not be of the same construction but the first and second circulators,  100  and  101 , are preferably of the same construction and are more preferably electrically matched. If dictated by physical constraints of the application, the first and second circulators,  100  and  101 , need not be physically identical, e.g., they may be mirror images or otherwise physically differ, but the first and second circulators,  100  and  101 , are preferably electrically similar in both amplitude and phase characteristics. 
     The transmitter output is connected to the input port  51  of the routing device  50 . The receiver input is connected to output port  53  of the routing device  50 . The antenna is connected to common port  52 . The transmission signal enters input port  51 , travels along transmission signal input connecting line  115  and enters an input port  116  of the input quadrature hybrid  110 . This signal that enters the input quadrature hybrid  110  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  118  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  117 . An isolated port  119  of the input quadrature hybrid  110  is terminated with a termination  120  in order to absorb reflected energy that may be entering port  118  and port  117 . The termination  120  will also absorb energy that leaks from the input port  116  to the isolated port  119 . 
     The first half of the signal derived from the transmission signal leaves port  118  of input quadrature hybrid  110 , propagates down connecting line  121  and enters port  123  of the first circulator  100 . Rotation of the first circulator  100  is shown as clockwise which implies that a signal entering port  123  will leave through port  125  of the first circulator  100 . This signal continues along connecting line  130  until it enters the port  131  of the common quadrature hybrid  111 . The common quadrature hybrid  111  is used for both transmitting signals and receiving signals through the routing device  50 . The signal entering port  131  of the common quadrature hybrid  111  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  135  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  134 . 
     The second half of the signal derived from the transmission signal leaves port  117  of input quadrature hybrid  110 , propagates down connecting line  122  and enters port  124  of the second circulator  101 . Rotation of the second circulator  101  is shown as counter-clockwise which implies that the signal entering the port  124  will leave through port  126 . This signal continues along feed line  132  and enters port  133  of the common quadrature hybrid  111 . The signal entering port  133  of the common quadrature hybrid  111  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  134  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  135 . 
     The equal amplitudes and relative phases of the two signals leaving port  134  of common quadrature hybrid  111  result in signal addition of the transmitted signal that enters the transmitter input port  51 . The transmitted signal travels down connecting line  137  and leaves the routing device  50  at the common port  52 . The equal amplitudes and relative phases of the two transmitted signals leaving port  135  of common quadrature hybrid  111  result in signal cancellation at the output port  135 . In the ideal case, no portion of the transmitted signal is absorbed in the termination  136  connected to the common quadrature hybrid  111  at port  135 . In the ideal case, the transmitted signal entering the input port  51  of the routing device  50  is first divided and then recombined to leave the routing device  50  at common port  52 . In practice, the total transmitted energy leaving port  52  will be reduced by the insertion loss of components and connecting lines used in routing device  50 . 
     The received signal entering the common port  52  of the routing device  50  travels along connecting line  137  and enters port  134  of common quadrature hybrid  111 . The signal that enters the common quadrature hybrid  111  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  133  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  131 . The isolated port  135  of the common quadrature hybrid  111  is terminated with a termination  136  in order to absorb any reflected received energy that may be entering port  131  and port  133 . The termination  136  will also absorb energy that leaks from the input port  134  to the isolated port  135 . 
     The first half of the received signal leaving port  133  travels along connecting line  132  and enters the second circulator  101  at the common port  126 . Rotation of the second circulator  101  is shown as counter-clockwise which implies that the signal entering the port  126  will leave through output port  128 . This signal continues along feed line  141  and enters port  143  of the output quadrature hybrid  112 . The signal entering port  143  of the output quadrature hybrid  112  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  145  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  144 . 
     The second half of the signal derived from the received signal leaves port  131  of common quadrature hybrid  111 , propagates down connecting line  130  and enters the common port  125  of the first circulator  100 . Rotation of the first circulator  100  is shown as clockwise which implies that the signal entering the common port  125  will leave through port  127 . This signal continues along feed line  140  and enters port  142  of the output quadrature hybrid  112 . The signal entering port  142  of the output quadrature hybrid  112  is split into two substantially equal amplitude signals with quadrature phase. One half of the signal input leaves port  144  with a relative phase of −90 degrees in relation to another half of the signal input that leaves through port  145 . 
     The equal amplitudes and relative phases of the two signals leaving port  144  of output quadrature hybrid  112  result in signal addition of the received signal. The received signal travels down connecting line  147  and leaves the routing device  50  at the receive port  53 . The equal amplitudes and relative phases of the two signals leaving port  145  of output quadrature hybrid  112  result in signal cancellation at the output port  145 . In the ideal case, no received signal entering the common port  52  of the routing device  50  is absorbed in the termination  146  connected to the output quadrature hybrid  112  at output port  145 . In the ideal case, the received signal entering the common port  52  of the routing device  50  is first divided by common quadrature hybrid  111  and then recombined by output quadrature hybrid  112  to leave the routing device  50  at receive port  53 . In practice the total received energy leaving receive port  53  will be reduced by the insertion loss of components and connecting lines used in routing device  50 . 
     It will be understood by those skilled in the art in view of this disclosure that the rotation of first circulator  100  and second circulator  101  in  FIG. 5  was chosen for clarity in the diagram and that the rotation direction of the first and second circulators,  100  and  101 , can be changed as long as the interconnecting lines are appropriately arranged to route the signals as described above. 
     The routing device  50  is designed to provide isolation between the transmit channel to the receive channel from any portion of the transmit signal that may couple through the first circulator  100  and second circulator  101 . In the ideal case, any signal entering the input port  123  will leave through common port  125  and no portion of the transmitted signal will be seen at output port  127 . In practice the first circulator  100  has limited amount of isolation between the input port  123  and output port  127 . This undesired coupling of energy from the input port  123  to the output port  127  is caused predominately by practical limitations in the circulator design and mismatch between common port  125  and connection to the connecting line  130 . The portion of the transmitted signal that couples through first circulator  100  will travel along connecting line  140  and enter the output quadrature hybrid  112  at the input port  142 . The coupled signal is split into two equal amplitude signals in quadrature phase. One half of the signal is delivered to the isolated port  145  and one half is delivered to the output port  144 . 
     In the ideal case any signal entering the input port  124  will leave through common port  126  and no portion of the transmitted signal will be seen at output port  128 . In practice the second circulator  101  has limited amount of isolation between the input port  124  and the output port  128 . This undesired coupling of energy from the input port  124  to the output port  128  is caused predominately by practical limitations in the circulator design and mismatch between common port  126  and connection to the connecting line  132 . The portion of the transmitted signal that couples through second circulator  101  will travel along connecting line  141  and enter the output quadrature hybrid  112  at the input port  143 . The coupled signal is split into two equal amplitude signals in quadrature phase. One half of the signal is delivered to the isolated port  145  and one half is delivered to the output port  144 . 
     It can be shown that undesired coupled signals through first circulator  100  and second circulator  101  will result in two equal amplitude signals appearing at the isolated port  145  and two equal amplitude signals at output port  144 . It can also be shown that the phase relationship between these signals will result in signal addition at the isolated port  145  and signal cancellation at output port  144 . In this way, any energy that is coupled through first circulator  100  and second circulator  101  will be terminated by termination  146  and no undesired coupled energy will be delivered to output port  144 . Output port  144  can be connected to the receive channel of a full duplex transceiver thus providing high isolation between the transmit channel to the receive channel. 
     As previously mentioned, it is expected that circulators  100  and  101  have approximately the same electrical performance in both amplitude and phase in order to maintain the quadrature phase relationship developed by the input quadrature hybrid  110 . Tuning elements and/or phase adjustment may be inserted along any feed line in order to adjust the amplitude and phase of the signal traveling along the line. Tuning the signal may improve the isolation between the transmit channel and receive channel by compensating for any differences between the signal paths. It is also found that tuning elements, such as small stubs, placed on connecting line  130  and/or connecting line  132  and placed in close proximity to circulator common port  125  and/or circulator common port  126  can greatly improve the amount of isolation between the transmit and receive channels. The tuning element or elements achieve a better electrical match between the two circulators. 
       FIG. 6  shows the routing device  50  for signals that travel from the transmitter to the first circulator  100  and the second circulator  101 . The complex input signal S 1  to the routing device  50  will be assumed to have voltage amplitude equal to 1 and phase equal to 0 degrees. TABLE 1 summarizes the amplitudes and relative phases for the signals traveling from the input port  51  up to the first and second circulators  100  and  101  respectively. As shown in  FIG. 6 , the input signal S 1  enters the input quadrature hybrid  110  at port  116  and the signal is divided into two equal amplitude signals with quadrature phase. The signal S 2  leaving port  118  has amplitude equal to 1/sqrt(2) and relative phase equal to −90 degrees and the signal S 3  leaving port  117  has amplitude equal to 1/sqrt(2) and relative phase equal to 0 degrees. The input quadrature hybrid  110  can also be configured with these two connections swapped. In this case, the connections to the other two quadrature hybrids would also need to be swapped in order to maintain the same performance. The two output signals from the first quadrature hybrid  110  travel along connecting lines  121  and  122  respectively. The length of transmission line for connecting lines  121  and  122  introduce an additional phase shift of −φ1 to each signal S 4  and S 5 . 
     
       
         
           
               
               
               
               
             
               
                   
                 TABLE 1 
               
               
                   
                   
               
               
                   
                 Signal 
                 Amplitude 
                 Phase 
               
               
                   
                   
               
             
            
               
                   
                 S1 
                 1 
                  0 
               
               
                   
                 S2 
                 1/sqrt(2) 
                 −90 
               
               
                   
                 S3 
                 1/sqrt(2) 
                  0 
               
               
                   
                 S4 
                 1/sqrt(2) 
                 −90 − φ1 
               
               
                   
                 S5 
                 1/sqrt(2) 
                 −φ1 
               
               
                   
                 S16 
                 B/sqrt(2) 
                 −90 − φ1 − φB 
               
               
                   
                 S17 
                 B/sqrt(2) 
                 −φ1 − φB 
               
               
                   
                 S18 
                 B 
                 −90 − φ1 − φB − φ3 
               
               
                   
                 S19 
                 0 
               
               
                   
                   
               
            
           
         
       
     
       FIG. 7  shows the signal paths for the coupled or leakage signals from the input port  123  and port  124  of circulators  100  and  101  respectively to the output ports  127  and  128 . The upper and lower sections of the routing device  50  are not shown for clarity. For this analysis, it is assumed that any undesired signal that couples through the circulator will experience a change in amplitude equal to B and a phase shift equal to −φB. The signal S 16  from the output port  127  will have an amplitude equal to B/sqrt(2) and relative phase of (− 90 −φ1−φB) degrees. The signal S 17  on the output port  128  will have amplitude equal to B/sqrt(2) and relative phase of (−φ−φB) degrees. These signals travel along feed lines  140  and  141  respectively. The length of transmission line for connecting lines  140  and  141  introduce an additional phase shift of −φ3 to each signal. Each input signal to the output quadrature hybrid  112  is divided in half. A relative phase shift of −90 degrees is introduced into the signal passing from the port  143  over to the port  145 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  142  over to the port  144 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  142  over to the port  145 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  143  over to the port  144 . Vector addition of the output signals at port  144  of the quadrature hybrid  112  will show signal cancellation resulting in output amplitude S 19  of 0. Vector addition of the output signals at port  145  of the quadrature hybrid  112  will show signal addition resulting in output amplitude S 18  of B. Output port  144  is connected to the receive channel to prevent undesired circulator coupling or leakage from entering the receiver. Port  145  is connected to termination  146  in order to terminate the undesired energy that coupled through the circulators. In some systems, the energy at the terminated port  145  can be measured and used as an indication of the operation of the circulators. For example, if a large signal level is measured at the port  145  then it may indicate a problem with the one or both circulators, as most of the signal is being coupled across the circulator and not properly transmitted through the antenna into the surrounding environment. 
     The above derivation assumed that the two signal paths were balanced in both relative amplitude and relative phase in order that signal cancellation would occur at the output port  144  of the routing device  50 . Tolerances in the components and connecting lines may result in a degradation of the transmit-to-receive isolation provided by the routing device  50 . A study of the amplitude balance and phase balance for the signals entering the quadrature hybrid  112  can show what level of transmit-to-receive isolation is achievable in the routing device  50 . Also note, that the quadrature hybrid  112  or other power combiner may also have relative amplitude and phase imbalance that may reduce the isolation performance. In this case, the tolerance within the quadrature hybrid  112  or other power combiner can be considered as part of the following analysis. TABLE 2 shows the required amplitude and phase balance between two signal paths that would result in a 30 dB and 40 dB isolation between the transmit channel to receive channel. TABLE 2 lists the required relative amplitude and phase tolerance as a function of the signal level of the undesired coupling. It is assumed that the amplitude and phase imbalances are created by differences in the insertion loss and electrical lengths of the connecting lines, electrical variations between the ports of the power dividers and combiners and electrical variations between the pair of circulators. For example, circulators that have a poor isolation such as 10 dB, would require tighter tolerance in the balance between the two combined signals in order to achieve a high isolation between transmit and receive channels. 
     As a numerical example using the TABLE 2, if the required transmit-to-receive isolation is 30 dB using the routing device  50  and the circulator isolation having a value of 15 dB, then the relative amplitude balance between the two paths would need to be within the range of +3.8 dB/−1.6 dB. This analysis assumes that the phase balance is ideal. Using this same example but with an ideal amplitude balance, the relative phase balance between the two paths would be +/−20.5 degrees. For the signal routing assembly having both amplitude and phase imbalances, a Monte Carlo analysis is one technique that can be used to estimate the range of tolerances required to achieve a certain level of isolation between the transmit channel to receive channel. For example, using a circulator with isolation of 10 dB would require a relative amplitude balance of +1.2 dB/−0.8 dB and a relative phase balance+/−10 degrees in order to achieve approximately 30 dB isolation between the transmit channel to receive channel. There are other combinations of amplitude and phase tolerances that can achieve this isolation value. 
     In practice, amplitude and phase adjustments within the signal routing assembly  50  can be implemented to improve the final isolation of the network. In this case, amplitude and phase shift tuning, using such components as attenuators and lengths of transmission lines, can adjust the balance between the two signal paths in order to optimize the isolation between the transmit channel and receive channel. In addition, proper selection of the components, and when using a printed circuit board, symmetrical layout of the connecting lines, can result in amplitude and phase balances within +/−0.3 dB and +/−5 degrees with minimal tuning at an operation frequency of 915 MHz. These tolerances can achieve approximately a 35 dB isolation between transmit to receive channels. 
     
       
         
           
               
               
               
             
               
                 TABLE 2 
               
               
                   
               
               
                 Undesired Signal 
                 Amplitude 
                 Phase 
               
               
                 Level (dB) 
                 Balance (dB) 
                 Balance (deg) 
               
               
                   
               
             
            
               
                   
               
            
           
           
               
            
               
                 Transmit to Receive Isolation = 30 dB 
               
            
           
           
               
               
               
            
               
                 5 
                     +1/−0.9 
                 +7−6.4 
               
               
                 10 
                 +1.9/−1.6 
                 +/−11.4 
               
               
                 15 
                 +3.8/−2.6 
                 +/−20.5 
               
               
                 20 
                 +8.7/−4.2 
                 +/−36.9 
               
               
                 25 
                     +inf/−6.5 
                 +/−68.4 
               
            
           
           
               
            
               
                 Transmit to Receive Isolation = 40 dB 
               
            
           
           
               
               
               
            
               
                 5 
                   +/−0.3 
                 +/−2.0  
               
               
                 10 
                  0.6/−0.5 
                 +/−3.6  
               
               
                 15 
                     +1/−0.9 
                 +/−6.4  
               
               
                 20 
                 +1.9/−1.6 
                 +/−11.4 
               
               
                 25 
                 +3.8/−2.6 
                 +/−20.5 
               
               
                   
               
            
           
         
       
     
     From the above analysis and data, it will be understood by those skilled in the art that amplitude levels that are exactly the same or phase differences that are exactly 180 degrees, while desirable for the practice of this invention, are not required for the practice of this invention. As indicated in the above Table 2, the amplitude balance and phase balance required to practice the invention will depend on the desired transmit to receive channel isolation and the undesired signal level produced by the transmission leakage through the circulators. The undesired signal levels are presented in terms of attenuation of the transmission input signal, i.e., the attenuation of the transmission signal passed from port one of the routing devices,  100  and  101 , which results in the undesired signal appearing at the combining assembly. Thus, for the present invention, the requirements for approximately the same level signals and approximately the desired phase shift, e.g., 180 degrees, are understood to mean within tolerances yielding a desired isolation based on the characteristics of the signal routing devices,  100  and  101 . Such tolerances are illustrated in the Table 2 for transmit to receive channel isolation levels of 30 dB and 40 dB. The undesired signal referred to is the leakage transmission signal from one of the routing devices,  100  and  101 , the value in dB represents the attenuation ratio relative to the divided transmission signals at the first ports of the routing devices,  100  and  101 , for the leakage transmission signal. 
     In practice the amount of cancellation in the signal combiner  112  varies with the matching of the signal. It is considered that the undesired leakage signals substantially cancel when the receiver front end functions adequately. Depending on the application, the amount of cancellation necessary will vary on the amount of leakage in the routing devices  100  and  101 . In applications such as RFID tag excitation and reading, it may be acceptable that the first and second leakage signals substantially cancel each other such that a signal appearing at the received signal output of the signal combiner  112  which is produced by the transmission signal, and does not include any signal received by the antenna by reception of radiation, is at least 20 dB below a level of the desired transmission signal. Preferably, such a signal is 25 dB down, more preferably such a signal is 30 dB down, and still more preferably such a signal is 40 dB down. It should further be noted that this cancellation is achieved routing the signals using passive components without employing active cancellation. 
       FIG. 8  shows two measured results for transmit channel to receive channel isolation. The upper curve  150  in  FIG. 8  is the isolation for the standard lumped element type circulator as shown in  FIG. 1 . The lumped element circulator was manufactured for best performance in the 902 MHz to 928 MHz frequency range. This measurement was made by measuring the difference in the leakage signal level  8  relative to the input signal  6  as shown in  FIG. 1 . A routing device was fabricated using a printed circuit etched onto a FR- 4  dielectric substrate. The routing device was fabricated with two lumped element circulators of the same type used in the first measurement  150  shown on  FIG. 8 . The circulators were manufactured for best performance in the 902 MHz to 928 MHz frequency range. The lower curve  151  in  FIG. 8  was measured using the preferred embodiment of routing device  50  as shown in  FIG. 5 . This measurement was made by measuring the signal level between receive channel output  53  relative to the signal level at the transmit channel input  51 . It is shown from the measured results that the routing device  50  provides a much lower isolation over a much wider range of frequencies. For example, the measured worst case isolation over the operating band of 860 MHz to 960 MHz is 13 dB for the standard lumped element circulator and 35 dB using the preferred embodiment for the routing device  50 . 
       FIG. 9  shows the measured results for the receive channel to transmit channel isolation. The upper curve  152  shows the measured isolation for the standard lumped element circulator as shown in  FIG. 1 . The standard circulator provides little isolation (&lt;1 dB) between the receive channel to transmit channel. The lower curve  153  is the measured isolation using the preferred embodiment of the routing device  50  as shown in  FIG. 5 . As shown in  FIG. 9 , the receive channel to transmit channel isolation is greater than 25 dB over the 860 MHz to 960 MHz frequency range. 
     Another embodiment of the present invention makes use of directional couplers in place of the circulators to route the signals to and from the common antenna port through the routing device  50 .  FIG. 10  shows the routing device  50  implemented with directional couplers  155  and  156 . The mathematical analysis using directional couplers in place of circulators follows the same derivation as shown in  FIG. 6  and  FIG. 7 . One of the key differences when using directional couplers in place of circulators is an additional reduction in the received amplitude of the signals as they pass through the directional coupler moving from connecting lines  130  and  132  to connecting lines  140  and  141  respectively as shown on  FIG. 10 . Also note that practical directional couplers have undesired leakage paths between the ports  157  to port  159  and port  160  to port  162 . As in the case using circulators, the routing device  50  is capable of canceling the undesired leakage energy at the output port  144  and allowing this energy to be absorbed in the termination  146 . 
     Another embodiment of the present invention replaces the quadrature hybrids  110 ,  111  and  112  in  FIG. 5  and  FIG. 10  with other types of power division networks as long as the output signals from these devices maintain the amplitude and the relative phase relationships required for proper operation of the routing device  50 . One skilled in the art will recognize in view of this disclosure other types of power dividers that have equal amplitude split with a 90-degree phase difference between the outputs that can be used to practice this invention such as the branchline coupler, overlay coupler, edge-coupled coupler, lumped element coupler and Lange coupler. Likewise, other types of power division networks with equal amplitude but equal phase between the outputs may be employed to practice the present invention. These equal phase dividers include the Wilkinson tee, resistive divider and T-junction or reactive tee. Using one of these equal amplitude-equal phase dividers in place of quadrature hybrid  110 ,  111  and/or  112  requires the addition of a 90-degree phase shift network on one side of the divider output. 
     For example,  FIG. 11  shows another embodiment of the routing device  50  using Wilkinson dividers  200 ,  201  and  202  in place of the three quadrature hybrids  110 ,  112  and  111  respectively as shown on  FIG. 5 . To create the required quadrature signal, additional 90-degree phase shifts  203 ,  204  and  205  are added to connecting lines  121 ,  140 ,  132  respectively to create the necessary conditions for isolating the transmit signal from entering the receive channel. The Wilkinson tee divider or any other type of equal phase power divider/combiner in combination with a 90-degree phase shift can also be used within the routing device  50 . The termination  206  is used to absorb the signals that leak or couple through circulators  100  and  101 . Additionally, it is realized that different combinations of divider types can be used in the routing device  50  to provide isolation between the transmit channel and receive channel. 
     One skilled in the art will understand in light of this disclosure that other types of power divider networks are usable in the practice of this invention that result in a variety of phase differences between the divider&#39;s output signals. For example, the ring hybrid, or “rat-race”, results in a power division with a 180-phase difference between two of the output ports. Here again, a phase shift network will be required to properly adjust the phase so that signals that leak or couple through the two circulators or directional couplers will be isolated from the receive channel. 
     The routing device of the present invention may also include amplifiers in the connecting lines to up to the circulators or directional couplers increase the amplitude level of the transmitted signal. These amplifiers should provide approximately an equal amount of amplification to the input signals and approximately an equal amount of phase shift. 
     The routing device of the present invention may also include modulators in the connecting lines to allow the routing device to operate as a transmit modulator as shown in  FIG. 12 . For example, modulators  224  and  225  are placed along connecting lines  121  and  122  respectively. Data signals are applied to the data input ports  226  and  227  and the transmission signals flowing on connecting lines  121  and  122  are modified by the modulators  224  and  225 . The modulators  224  and  225  can be mixers, switches, variable attenuators, variable amplifiers or any device that can modify the amplitude and/or phase of the transmission signal. In the typical operation of an RFID system using backscatter communication, the reader modulation is applied during forward-link transmission from the RFID reader to the RFID tag. During reverse-link communication, the RFID reader transmitter is active but typically not modulated with data during signal reception from the tag to the reader. In this case, the routing device  220  provides isolation between the active transmitter carrier signal and receiver input. The routing device  220  may also include amplifiers in the connecting lines to increase the amplitude level of the transmitted signal to operative levels as shown in  FIG. 12 . For example, amplifiers  222  and  223  are placed along connecting lines  121  and  122  respectively. 
     The routing device of the present invention is optionally operated in full duplex mode with different transmit and receive RF carrier frequencies. In this way, cancellation of the transmit energy at frequency f 1  will be performed by the routing device allowing the receiver to be simultaneously receiving signals at a different frequency f 2 . The only limitation to the frequency spacing between f 1  and f 2  is the operational bandwidth of the circulators, couplers and dividing components used in the routing device. 
     It will also be appreciated in view of this disclosure that practical limitations in the performance of the divider/combiner  57  may introduce undesired leakage signals that may reduce the transmit to receive isolation of the signal routing device  50 . For example, a transmission leakage signal may exist in the divider/combiner  57  between ports  64  and  65 . A portion of the first divided transmission output signal entering port  64  of the divider/combiner  57  may undesirably leak to port  65  and appear at the output port  53  of the routing device  50 . This leakage is created by but not limited to the isolation of the divider/combiner  57  and reflection introduced at the connection between port  66  of the divider/combiner  57  and connecting line  74 . In the similar way, a portion of the second divided transmission output signal entering port  65  of the divider/combiner  57  may undesirably leak to port  64  and appear at the output port  53  of the routing device  50 . These leakage signals combine into a divider/combiner leakage signal, Ls, appearing at output port  53  of the routing device  50  and may interfere with the proper operation of the receiver. The divider/combiner leakage signal, Ls, is a complex value having an amplitude and relative phase. 
     It can be shown that when divider/combiner  57  has finite isolation between port  64  and port  65  then the transmit-to-receive isolation of routing device  50  will degrade. In practice, when divider/combiner  57  is a quadrature hybrid, coupled line coupler, branchline coupler, Lange coupler, rat race, ring hybrid or equal phase power combiner such as Wilkinson tee, resistive divider and T-junction or reactive tee, the port-to-port isolation will be in the range of 15-30 dB. The finite isolation creates a transmitter leakage signal that is not cancelled by the routing device  50 . In this case, the divider/combiner leakage signal, Ls, appearing at output port  53  of the routing device  50  is limited by the value of the finite isolation of the divider/combiner  57 . 
       FIG. 13  shows the signal paths for the leakage signals from port  131  to port  133  and from port  133  to port  131 . The upper and lower sections of the routing device  50  are not shown for clarity. For the transmitted signal entering the signal routing device  50  and divided into the first and second divided transmission signals and routed by the first and second routing devices  100  and  101  to the first and second divided transmission output signals represented in  FIG. 13  as S 20  and S 25  respectively and having substantially equal amplitudes and a relative phase shift therebetween. The first and second routing devices  100  and  101  are shown as circulators but can be any other routing device such as directional couplers or other routing device. For this analysis, the first and second divided transmission output signals S 20  and S 25  will be assumed to have a voltage amplitude of 1/sqrt(2) and relative phase difference equal to 90 degrees as listed in TABLE 3. For this analysis, it is assumed that the first and second routing devices  100  and  101  are ideal and signals entering port  123  and  124  are routed to ports  125  and  126  respectively with no change in amplitude and phase shift equal to −φ1. Also, signals entering ports  125  and  126  are routed to ports  127  and  128  with no change in amplitude and phase shift equal to −φ1. For this analysis, it is further assumed that the connecting lines  130  and  132  will introduce a phase shift of −φ4 degrees and that the connecting lines  140  and  141  will introduce a phase shift of −φ5 degrees. In practice, these connecting lines will have an associated insertion loss but the insertion loss will not be included as part of this analysis. A portion of second divided transmission output signal S 25  entering port  133  of common hybrid  111  will leak to port  131  as a third transmission leakage signal S 26 . A portion of the first divided transmission output signal S 20  entering port  131  of common hybrid  111  will leak to port  133  as a fourth transmission leakage signal S 21 . These undesired leakage signals are found in practice and not limited to quadrature hybrids. This leakage would also be present in coupled line coupler, branchline coupler, Lange coupler, rat race, ring hybrid or equal phase power combiner such as Wilkinson tee, resistive divider and T-junction or reactive tee. The leakage signal can be measured and/or calculated using standard techniques known in the industry. 
     For this analysis, the third and fourth transmission leakage signals S 26  and S 21  will experience a change in amplitude equal to H and a phase shift equal to −φH. The third and fourth transmission leakage signals S 26  and S 21  will travel along feed lines  130  and  132  respectively and be routed through the signal routing devices  100  and  101  respectively and exit through port  127  and  128  respectively. In practice, these transmission paths will include insertion loss and the amplitude and phase will be a function of frequency. These modified third and fourth transmission leakage signals entering port  142  and port  143  are represented as S 27  and S 22  respectively in  FIG. 13  and TABLE 3. The signal S 22  will have an amplitude equal to H/sqrt(2) and relative phase of (−90−2(φ1)−2(φ4)−φ5−φH) degrees. The signal S 27  will have amplitude equal to H/sqrt(2) and relative phase of (−2(φ1)−2(φ4)−φ5−φH) degrees. The power in each input signal to the output quadrature hybrid  112  is divided in half or the voltage amplitude is scaled by a factor of 1/sqrt(2) in voltage. A relative phase shift of −90 degrees is introduced into the signal passing from the port  143  over to the port  145 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  142  over to the port  144 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  142  over to the port  145 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  143  over to the port  144 . 
     Vector addition of these leakage signals at port  145  of the quadrature hybrid  112  will show signal cancellation resulting in output amplitude S 24  equal to 0. Vector addition of these leakage signals at port  144  of the quadrature hybrid  112  will show signal addition resulting in output amplitude S 23  equal to H and relative phase shift of (−90−2(φ1)−2(φ4)−φ5−φH). Output port  144  is connected to the receiver and the total leakage signal S 23  is undesired and may affect the proper operation of the receiver. The total leakage signal S 23  described here was previously referred to as divider/combiner leakage signal, Ls. Therefore, the divider/combiner leakage signal, Ls, will have a relative amplitude of H and a relative phase shift of (−90−2(φ1)−2(φ4)−φ5−φH).
 
 Ls=|Ls|∠φ   Ls   =H ∠(−90−2(φ1)−2(φ4)−φ5 −φH )
 
It is therefore necessary to eliminate or reduce the amplitude of the divider/combiner leakage signal, Ls, to an acceptable level for proper operation of the receiver.
 
     
       
         
           
               
               
               
               
             
               
                   
                 TABLE 3 
               
               
                   
                   
               
               
                   
                 Signal 
                 Amplitude 
                 Phase 
               
               
                   
                   
               
             
            
               
                   
                 S20 
                 1/sqrt(2) 
                 −90 − φ1 − φ4 
               
               
                   
                 S25 
                 1/sqrt(2) 
                 −φ1 − φ4 
               
               
                   
                 S21 
                 H/sqrt(2) 
                 −90 − φ1 − φ4 − φH 
               
               
                   
                 S26 
                 H/sqrt(2) 
                 −φ1 − φ4 − φH 
               
               
                   
                 S22 
                 H/sqrt(2) 
                 −90 − 2(φ1) − 2(φ4) − φ5 − φH 
               
               
                   
                 S27 
                 H/sqrt(2) 
                 −2(φ1) − 2(φ4) − φ5 − φH 
               
               
                   
                 S24 
                 0 
               
               
                   
                 S23 
                 H 
                 −90 − 2(φ1) − 2(φ4) − φ5 − φH 
               
               
                   
                   
               
            
           
         
       
     
     It was previously discussed that tuning elements and/or phase adjustment may be inserted along any connecting line in order to balance the amplitude and phase of the signals traveling within the signal routing device  50 . Unfortunately, tuning elements that “balance” or match signal paths will not reduce the amplitude of the divider/combiner leakage signal, Ls. In contrast to using tuning elements to balance the amplitude and phase characteristics, the present invention optionally provides for the use of reflector devices introduced to the routing device  50  and configured to “imbalance” a portion of the signal paths in order to introduce a compensating signal, Cs, that is substantially equal in amplitude to the divider/combiner leakage signal, Ls, but having approximately 180-degree relative phase difference for the purpose of reducing the amplitude of the divider/combiner leakage signal, Ls, appearing at output port  53  of the routing device  50 . In practice exact matching of Cs and Ls so as to be equal in amplitude and have exactly 180 degree phase difference is impracticable, hence the present invention is directed to an embodiment where this matching is substantially or approximately achieved such that the divider/combiner leakage signal, Ls, is reduced to a level permitting desired system operation such as or better than that illustrated in  FIG. 8  isolation characteristic  151 . Such reflector devices may include stubs or lumped components or other devices as are known by those skilled in the art. 
     The present invention further includes embodiments which include a reflector device to create an imbalance in the routing device  50  resulting in a compensation signal, Cs, that will effectively reduce the divider/combiner leakage signal, Ls, created from the finite isolation of the common quadrature hybrid  111 . 
     As described above and shown in TABLE 3, one of the limitations for achieving high transmit to receive isolation using the signal routing device  50  is the direct result of divider/combiner leakage signal, Ls. In order to reduce the effect of this leakage signal and improve the overall isolation of the signal routing device  50 , a separate compensating signal, Cs, can be added at the receiver port  53 . This additional compensating signal needs to have approximately the same amplitude as the divider/combiner leakage signal, Ls, and approximately 180-degree relative phase shift to the phase of the leakage signal. 
     The present invention provides for a reflector device  170  or  170 ′ respectively placed along connecting line  130  or  132  which will introduce an imbalance in routing device  50  and an associated compensating signal, Cs, that appears at the receiver port thus effecting an improvement in transmit to receive isolation when the compensating signal, Cs, is properly set to cancel the divider/combiner leakage signal, Ls. The present invention further provides a configuration wherein both reflector devices  170  and  170 ′ are used. In configurations when both reflector devices  170  and  170 ′ are simultaneously used, the combination can be set so they effect an imbalance in routing device  50  and the combined compensation signal, Cs, can also be used to effect a cancellation of the divider/combiner leakage signal, Ls.  FIG. 14  shows the signal paths for the signal S 30  reflected from reflector device  170  placed along connecting line  130 . The upper and lower sections of the routing device  50  are not shown for clarity. The reflected signal S 30  is a portion of the first divided transmission output signal leaving port  125  of the routing device  100 . For this analysis, the reflected signal S 30  entering port  125  of the first routing device  100  is assumed to have an amplitude X/sqrt(2) and relative phase (−φX−90−2(φ6)−φ1) degrees, as shown in TABLE 4, where the amplitude of the reflection from reflector device  170  is X and the relative phase of the reflection from reflector device  170  is −φX. It is also assumed that signals entering port  125  of signal routing device  100  is routed to port  127  with no change in amplitude and phase shift equal to −φ1 degrees. The portion of connecting line  130  between reflector device  170  and port  125  of routing device  100  will introduce a relative phase shift of −φ6 degrees. The length of connecting line  140  will introduce an additional phase shift of −φ5 degrees. The signal S 31  entering port  142  of output quadrature hybrid  112  will have an amplitude of X/sqrt(2) and relative phase of (−φX−90−2(φ6)−2(φ1)−φ5) degrees. The output quadrature hybrid  112  divides the input power to any port in half or the voltage is scaled by a factor of 1/sqrt(2). A relative phase shift of 0 degrees is introduced into the signal passing from the port  142  over to the port  145 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  142  over to the port  144 . The resulting signal S 32  leaving port  145  of output quadrature hybrid  112  will have an amplitude of X/ 2  and relative phase of (−φX−90−2(φ1)−2(φ1)−φ5) degrees. The resulting signal S 33  leaving port  144  of output quadrature hybrid  112  will have an amplitude of X/2 and relative phase of (−φX−180−2(φ1)−2(φ1)−(φ5) degrees. Reflected signal S 33  was previously referred to as compensating signal, Cs.
 
 Cs=|Cs|∠φ   Cs =( X /2)∠(−φ X −180−2(φ6)−2(φ1)−φ5−φ5)
 
The reflector device  170  and placement along connecting line  130  is set to provide a compensating signal, Cs, that is substantially equal in the amplitude to the divider/combiner leakage signal, Ls, and relative phase of approximately 180-degrees out of phase with the divider/combiner leakage signal, Ls. The vector addition of these signals will reduce or eliminate the divider/combiner leakage signal, Ls, thus improving the transmitter to receiver channel isolation.
 
| Cs|∠φ   Cs   =|Ls |∠(φ Ls −180)
 
for the amplitudes
 
| Cs|≈|Ls| 
 
( X/ 2)≈ H  
 
then
 
 X ≈(2 H )
 
for the phase,
 
∠φ Cs ≈(φ Ls −180)
 
(−φ X− 2(φ6)−2(φ1)−φ5−180)≈((−90−2(φ1)−2(φ4)−φ 5 −φ H )−180)
 
(−φ X− 2(φ6)≈((−90−2(φ4)−φ H )
 
then
 
−φ X ≈(−90−2(φ4)−φ H+ 2(φ6))
 
     As a result, the amplitude, X, of the reflected signal from reflector device  170  should be set to be substantially equal to twice the amplitude, H, of the leakage signal of common quadrature hybrid  111 . The relative phase, −φX, of the reflected signal from reflector device  170  should be set to be approximately equal to the (−90−2(φ4)−φH+2(φ6)) degrees where −φH is the phase shift of the leakage signal of common quadrature hybrid  111 . 
     
       
         
           
               
               
               
               
             
               
                   
                 TABLE 4 
               
               
                   
                   
               
               
                   
                 Signal 
                 Amplitude 
                 Phase 
               
               
                   
                   
               
             
            
               
                   
                 S30 
                 X/sqrt(2) 
                 −φX − 90 − 2(φ6) − φ1 
               
               
                   
                 S31 
                 X/sqrt(2) 
                 −φX − 90 − 2(φ6) − 2(φ1) − φ5 
               
               
                   
                 S32 
                 X/2 
                 −φX − 90 − 2(φ6) − 2(φ1) − φ5 
               
               
                   
                 S33 
                 X/2 
                 −φX − 180 − 2(φ6) − 2(φ1) − φ5 
               
               
                   
                   
               
            
           
         
       
     
     Reflector device  170  should be set to effect cancellation of the divider/combiner leakage signal, Ls, such that a transmit to receive isolation of at least 30 dB is achieved over a frequency range associated with the system use. More preferably, reflector device  170  should be set to effect leakage cancellation such that at least 35 dB isolation is achieved over the desired frequency range. Still more preferably, reflector device  170  should be set to effect leakage cancellation such that at least 40 dB isolation is achieved over the desired frequency range. 
     In the preferred embodiment of this invention, reflector device  170  and/or  170 ′ is an open stub transmission line.  FIG. 16A  shows a top view of the preferred embodiment using transmission line  180  that is a portion of one of the connecting lines previously described. Open circuit  182  is at the end of transmission line stub  181 . The length and width of transmission line stub  181  is set to effect cancellation of the divider/combiner leakage signal, Ls. Alternatively, the reflector device  170  and/or  170 ′ can be a shorted stub transmission line.  FIG. 16B  shows a top view of an embodiment using transmission line  180  with a short circuit  184  placed along transmission line stub  183 . The length and width of transmission line stub  183  is set to effect cancellation of the divider/combiner leakage signal, Ls. Alternatively, the reflector device  170  and/or  170 ′ can a lumped element type reactive component such as a capacitor or inductor.  FIG. 16C  shows a top view of an embodiment using transmission line  180  with a short circuit  187  placed at the end of reactive component  186 . It will be understood that  FIGS. 16A-16C  are not to scale and that they are schematic in nature and that actual implementation is dependent upon the materials and frequencies involved. Reactive component  186  is connected to transmission line stub  185 . The capacitance or inductance value of reactive component  186  and the length and width of transmission line stub  185  are set to effect cancellation of the divider/combiner leakage signal, Ls. 
     A similar mathematical derivation to that described above can show that a compensating signal reflected from a reflector device  170 ′ place along connecting line  132  will effect cancellation of the divider/combiner leakage signal, Ls. For this analysis, the reflected signal entering port  126  of the second routing device  101  is assumed to have an amplitude Y/sqrt(2) and relative phase (−φY−2(φ7)−φ1) degrees where the amplitude of the reflection from reflector device  170 ′ is Y and the relative phase of the reflection from reflector device  170  is −φY. It is also assumed that signals entering port  126  of signal routing device  101  is routed to port  128  with no change in amplitude and phase shift equal to −φ1 degrees. The portion of connecting line  132  between reflector device  170 ′ and port  126  of routing device  101  will introduce a relative phase shift of −φ7 degrees. The length of connecting line  141  will introduce an additional phase shift of −φ5 degrees. The signal entering port  143  of output quadrature hybrid  112  will have an amplitude of Y/sqrt(2) and relative phase of (−φY−2(φ7)−2(φ1)−φ5) degrees. The output quadrature hybrid  112  divides the input power to any port in half or the voltage is scaled by a factor of 1/sqrt(2). A relative phase shift of 0 degrees is introduced into the signal passing from the port  143  over to the port  144 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  143  over to the port  145 . The resulting signal leaving port  145  of output quadrature hybrid  112  will have an amplitude of Y/2 and relative phase of (−φY−2(φ7)−2(φ1)−(φ5−90) degrees. The resulting signal leaving port  144  of output quadrature hybrid  112  will have an amplitude of Y/2 and relative phase of (−φY−2(φ7)−2(φ1)−φ5) degrees. This reflected signal was previously referred to as compensating signal, Cs. 
     As a result, the amplitude, Y, of the reflected signal from reflector device  170 ′ should be set to be substantially equal to twice the amplitude, H, of the leakage signal of common quadrature hybrid  111 . The relative phase, −φY, of the reflected signal from reflector device  170 ′ should be set to be approximately equal to the (−270−φH−2(φ4)+2(φ7)) degrees where −φH is the phase shift of the leakage signal of common quadrature hybrid  111 . 
     It is important to note that it may be possible to effect cancellation of the divider/combiner leakage signal, Ls, with the introduction of two or more reflector devices placed along connecting line  130  and/or connecting line  132  and therefore effecting an imbalance in the routing device  50  for effecting cancellation of the divider/combiner leakage signal, Ls. 
     Another technique for reducing the amplitude of divider/combiner leakage signal, Ls, is provided by reflector device  171  placed along connecting line  137  between common port  134  of the common quadrature hybrid  111  and the common port  52  of the signal routing device  50  as shown in  FIG. 15 . Portions of the upper and the lower sections of routing device  50  in  FIG. 15  are not shown for clarity. Reflector device  171  reflects a portion of the transmission output signal back into common port  134  of common quadrature hybrid  111 . A substantial portion of this reflected signal will appear at port  144  of output quadrature hybrid  112  and may be used to reduce the amplitude of divider/combiner leakage signal, Ls. Reflector device  171  is set to effect an amplitude for the compensating signal, Cs′, appearing at port  144 , to be substantially equal in amplitude to the divider/combiner leakage signal, Ls, appearing at port  144 , and approximately 180-degrees in relative phase to the divider/combiner leakage signal, Ls, appearing at port  144 . For this configuration, the compensating signal will be referred with variable Cs′. The vector addition of divider/combiner leakage signal, Ls, and the compensating signal, Cs′, created from signal reflection from the reflector device  171  will result in signal cancellation at port  144  and improve the transmitter to receiver channel isolation.  FIG. 15  shows the signal path from the reflector device  171  to port  144  of the output quadrature hybrid  112 . TABLE 5 shows the amplitude and relative phase for the associated signals. For this analysis, it is assumed that signals entering port  125  and  126  of signal routing device  100  and  101  are routed to ports  127  and  128  respectively with no change in amplitude and phase shift equal to −φ1. The portion of connecting line  137  between reflector device  171  and common port  134  of common quadrature hybrid  111  will introduce a relative phase shift of −φ8 degrees. The signal S 34  is the reflected signal from the reflector device  171  and enters common port  134  of common quadrature hybrid  111 . For this analysis, the reflected signal S 34  is assumed to have an amplitude R and relative phase (−φR−φ1−φ4−2(φ8)−90) where the amplitude of the reflection from reflector device  171  is R and the relative phase of the reflection from reflector device  171  is −φR. The signal S 34  enters common quadrature hybrid  111  at port  134  and divided between port  131  and port  133 . The signal S 35  leaving port  131  will have an amplitude equal to R/sqrt(2) and relative phase of (−φR−φ1−φ4−2(φ1)−90) degrees. The signal S 36  leaving port  133  will have an amplitude equal to R/sqrt(2) and relative phase of (−φR−φ1−φ4−2(φ1)−180) degrees. The connecting lines  130  and  132  will introduce an additional phase shift of −φ4 degrees to each respective signal. The signal routing device  100  and  101  will introduce a relative phase shift of −φ1 degrees. The connecting lines  140  and  141  will introduce a phase shift of −φ5 degrees to each respective signal. The signal S 37  entering port  142  of output quadrature hybrid  112  will have a magnitude of R/sqrt(2) and relative phase of (−φR−2(φ1)−2(φ4)−2(φ8)−90−φ5) degrees. The signal S 38  entering port  143  of output quadrature hybrid  112  will have an amplitude of R/sqrt(2) and relative phase of (−φR−2(φ1)−2(φ4)−2(φ8)−180−φ5) degrees. The power in each input signal to the output quadrature hybrid  112  is divided in half or the voltage is scaled by a factor of 1/sqrt(2). A relative phase shift of −90 degrees is introduced into the signal passing from the port  143  over to the port  145 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  142  over to the port  144 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  142  over to the port  145 . A relative phase shift of 0 degrees is introduced into the signal passing from the port  143  over to the port  144 . 
     Vector addition of the reflected signals at port  145  of the output quadrature hybrid  112  will show signal cancellation resulting in a signal amplitude of signal S 39  equal to 0. Vector addition of the reflected signals at port  144  of the output quadrature hybrid  112  will show signal addition resulting in output amplitude of signal S 40  equal to R and relative phase (−φR−2(φ1)−2(φ1)−2(φ1)−180−φ5) degrees. Reflected signal S 40  is referred to as the compensating signal, Cs′.
 
 Cs′=|Cs′|∠φ   Cs′   =R ∠(−φ R −2(φ1)−2(φ4)−2(φ8)−180−φ5)
 
The reflector device  171  and placement along connecting line  137  is set to provide a compensating signal, Cs′, that is substantially equal in the amplitude to the divider/combiner leakage signal, Ls, and relative phase of approximately 180-degrees with the divider/combiner leakage signal, Ls. The vector addition of these signals will reduce or eliminate the amplitude of the divider/combiner leakage signal, Ls, thus improving the transmitter to receiver channel isolation.
 
| Cs′|∠φ   Cs′   =|Ls |∠(φ Ls −180)
 
for the amplitudes
 
 |Cs′|≈|Ls| 
 
then
 
 R≈H  
 
for the phase,
 
∠φ Cs′ ≈∠(φ Ls −180)
 
(−φ R− 2(φ1)−2(φ4)−2(φ8)−180−φ5)≈((−90−2(φ1)−2(φ4)−φ5−φ H )−180)
 
then
 
−φ R≈−φH −90+2(φ8)
 
     As a result, the amplitude, R, of the reflected signal from reflector device  171  should be set to be substantially equal the amplitude, H, of the leakage signal of common quadrature hybrid  111 . The relative phase, −φR, of the reflected signal from reflector device  171  should be set to be approximately equal to the (−φH−90+2(φ8)) degrees where −φH is the phase shift of the leakage signal of common quadrature hybrid  111 . The relative phase is a modulo function of 360-degrees so that the relative phase, −φR, of the reflected signal from reflector device  171  can also be set to be approximately equal to the (−(φH−90+2(φ8)−n(360)) degrees, where n= . . . , −2, −1, 0, 1, 2, 3, 4, . . . Reflector device  171  should be set to effect cancellation of the divider/combiner leakage signal, Ls, such that a transmit to receive isolation of at least 30 dB is achieved over a frequency range associated with the system use. More preferably, reflector device  171  should be set to effect leakage cancellation such that at least 35 dB isolation is achieved over the desired frequency range. Still more preferably, reflector device  171  should be set to effect leakage cancellation such that at least 40 dB isolation is achieved over the desired frequency range. 
     In the preferred embodiment of this invention, reflector device  171  is an open stub transmission line.  FIG. 16A  shows a top view of the preferred embodiment using transmission line  180  that is a portion of one of the connecting lines previously described. Open circuit  182  is at the end of transmission line stub  181 . The length and width of transmission line stub  181  is set to effect cancellation of the divider/combiner leakage signal, Ls. Alternatively, the reflector device  171  can be a shorted stub transmission line.  FIG. 16B  shows a top view of an embodiment using transmission line  180  with a short circuit  184  placed along transmission line stub  183 . The length and width of transmission line stub  183  is set to effect cancellation of the divider/combiner leakage signal, Ls. Alternatively, the reflector device  171  can a lumped element type reactive component such as a capacitor or inductor.  FIG. 16C  shows a top view of an embodiment using transmission line  180  with a short circuit  187  placed at the end of reactive component  186 . Reactive component  186  is connected to transmission line stub  185 . The capacitance or inductance value of reactive component  186  and the length and width of transmission line stub  185  are set to effect cancellation of the divider/combiner leakage signal, Ls. 
     
       
         
           
               
               
               
             
               
                 TABLE 5 
               
               
                   
               
               
                 Signal 
                 Amplitude 
                 Phase 
               
               
                   
               
             
            
               
                 S34 
                 R 
                 −φR − φ1 − φ4 − 2(φ8) − 90 
               
               
                 S35 
                 R/sqrt(2) 
                 −φR − φ1 − φ4 − 2(φ8) − 90 
               
               
                 S36 
                 R/sqrt(2) 
                 −φR − φ1 − φ4 − 2(φ8) − 180 
               
               
                 S37 
                 R/sqrt(2) 
                 −φR − 2(φ1) − 2(φ4) − 2(φ8) − 90 − φ5 
               
               
                 S38 
                 R/sqrt(2) 
                 −φR − 2(φ1) − 2(φ4) − 2(φ8) − 180 − φ5 
               
               
                 S39 
                 0 
               
               
                 S40 
                 R 
                 −φR − 2(φ1) − 2(φ4) − 2(φ8) − 180 − φ5 
               
               
                   
               
            
           
         
       
     
     It will be appreciated that a reflector device introduced to create a compensating signal, Cs or Cs′, to effect cancellation of the amplitude of the divider/combiner leakage signal, Ls, can also be implemented in routing device  50  when directional couplers  155  and  156  are used in place of circulators  100  and  101 . As shown in  FIG. 10 , leakage from common quadrature hybrid  111  is still present in this configuration and any divider/combiner leakage signal found on connecting line  147  can be cancelled through the use of a reflector device placed on connecting line  137  and/or connecting line  130  and/or connecting line  132 . 
     Another embodiment of a divider/combiner leakage cancellation configuration includes a magnetic biasing device applied to one of the circulators so as to effect an imbalance in the first and second transmission leakage signals and third and fourth transmission leakage output signals resulting in cancellation of the first and second transmission leakage signals and third and fourth transmission leakage output signals at the received signal output. The magnetic biasing device is a magnetic device, a metallic device or a coiled wire carrying electrical current that is placed in the vicinity of the circulator. The basic construction of the circulator contains a ferrite material that is internally biased with a magnet. The operation and performance of the circulator is directly related to the interaction of the magnetic bias with the ferrite material. The introduction of an external magnetic bias placed above, below or at the sides of the circulator will result in change in the total magnetic bias and tuning of the circulator.  FIG. 17  shows a configuration for placing a magnetic biasing device  400  over circulator  100 .  FIG. 17  also shows a configuration for placing a magnetic biasing device  401  over circulator  101 . Applying one of biasing device  400  or  401  will result in an imbalance for the modified third and fourth transmission leakage output signals S 27  and S 22  respectively and first and second transmission leakage signals S 16  and S 17 , shown in  FIG. 7 , respectively and will achieve cancellation of the first and second leakage transmission leakage signals and the third and fourth transmission leakage output signals at the received signal output leaving port  144 . 
     The magnetic biasing device can be a neodymium magnet, alnico magnetic, steel or other material that is capable of affecting the total magnetic bias of the circulator. For example, an ultra-high-pull neodymium disc magnet placed above the circulator will achieve cancellation of the first and second leakage transmission leakage signals and the third and fourth transmission leakage output signals at the received signal output. The physical size of the magnetic biasing device can be larger or smaller than the actual circulator. For example, a lumped element circulator with physical size of 5 mm by 5 mm by 2 mm in height may be tuned with a neodymium magnet with diameter 2 mm and height 1.5 mm and positioned on top of the circulator. Tuning may occur while observing the measured isolation of routing device  50  and the magnetic biasing device  400  is moved around circulator  100  or magnetic biasing device  401  is moved around circulator  101  until the desired isolation is achieved. 
     Additionally, changing the magnetic bias of the circulator may be accomplished with a coil of wire carrying an electrical current. When a coil of wire has an applied current, a magnetic field develops through the center of the coil along its longitudinal axis. The magnetic field increases with larger current and with increasing the number of loops in the coil. When placed in the vicinity of the circulator, the coil&#39;s magnetic field will change the total magnetic bias of the circulator and will achieve cancellation of the first and second leakage transmission leakage signals and the third and fourth transmission leakage output signals at the received signal output 
     In addition, changing the magnetic biasing of the circulator may be performed with physical or electrical changes to the internal biasing and tuning of the circulator itself. Tuning the internal magnetic bias of the circulator may be accomplished during the construction of the circulator by adjusting the position of the circulator&#39;s internal components or housing. A typical circulator construction includes a number of layers comprised of ferrites, magnets, pole pieces, ground plates and a center conductor. These layers often referred to as a “stack” and can be adjusted or tuned before fixed in place either by mechanical means or soldered together. Tuning is also accomplished by shaping the center conductor of the stack or with quarter wave transformers or open-ended tuning stubs positioned around the center conductor of the stack. Additionally, tuning the internal magnetic bias of the circulator may be accomplished after construction by adjusting the position of the circulator&#39;s internal components or housing by means of applied mechanical pressure. Additionally, circulator construction with internal components fixed in place with soldered connections allow tuning by reflowing the solder of the housing and applying an external pressure on the lid of the housing while the solder is in a liquid state Changing the magnetic bias of one circulator will imbalance the first and second leakage transmission leakage signals and third and fourth transmission leakage output signals resulting in a cancellation of the first and second leakage transmission leakage signals and third and fourth transmission leakage output signals at the received signal output. 
     Further, the signal routing assembly  50  of  FIG. 17  is understood to be an embodiment of a portion of the generalized signal routing assembly  50  of  FIG. 4  which optionally includes elements of the embodiment of  FIG. 5  which in turn include the transmission signal input  51  for receiving the first transmission signal, the common port  52  for outputting the portion of the first transmission signal from transmission signal input and, when present, receiving the second transmission signal, the transmission signal output  53  for outputting the portion of the second transmission signal when present. Signal routing assembly  50  also includes the signal divider  110  receiving the first transmission signal from the transmission signal input and dividing the first transmission signal into the first and second divided transmission signals having amplitudes within the first amplitude range of each other and the first relative phase shift. The first routing device  100  and second routing device  101  each having at first, second and third ports, with the first and second routing devices being configured to simultaneously deliver the signal at the first port to the second port and another signal at the second port to the third port where the following transmission coefficients are defined as
         s 21  is a transmission coefficient from the first port to the second port;   s 32  is a transmission coefficient from the second port to the third;   s 31  is a transmission coefficient from the first port to the third port;   s 21  is greater than s 31 ; and s 32  is greater than s 31 ;       

     The first routing device  100  having the first divided transmission signal applied to the first port  123  of the first routing device  50  and simultaneously producing the first divided transmission output signal at the second port  125  of the first routing device  50  and the first transmission leakage signal at the third port  127  of the first routing device  100 , each the resultant from the first divided transmission signal. Also the first routing device  100  also has the third divided transmission signal, when present, applied to the second port  125  of the first routing device  100  and producing, simultaneously with the first divided transmission output signal and the first transmission leakage signal, the third divided transmission output signal at the third port  127  of the first routing device  100  resulting from the third divided transmission signal. 
     The second routing device  101  having the second divided transmission signal applied to the first port  124  of the second routing device  101  and simultaneously producing the second divided transmission output signal at the second port  126  of the second routing device  101  and the second transmission leakage signal at the third port  128  of the second routing device  101  each resulting from the second divided transmission signal. The second routing device  101  also having the fourth divided transmission signal, when present, applied the second port  126  of the second routing device and producing, simultaneously with the second divided transmission output signal and the second transmission leakage signal, the fourth divided transmission output signal at the third port  128  of the second routing device  101  resulting from the fourth divided transmission signal. 
     The signal routing assembly  50  of  FIG. 17  also contains the signal divider/combiner  111  having the first divider/combiner port  131  and the second divider/combiner port  133  for receiving the first and second divided transmission output signals and the configuration for combining these signals at levels within the second relative amplitude range and with the second relative phase shift to output the portion of the first transmission signal to the common port  52 . This configuration being such that the second transmission signal, when present and received at the common port  52 , that the second transmission signal is divided into the third and fourth divided transmission signals having amplitudes within the second relative amplitude range and with the second relative phase shift. 
     The signal routing assembly  50  of  FIG. 17  also contains the signal combiner  112  having first combiner input  142  and second combiner input  143  for receiving the first and second transmission leakage signals and, when present, for receiving the third and fourth divided transmission output signals. The signal combiner  112  includes the transmission signal output  53  and the signal combiner  112  and is configured to introduce the third relative phase shift into signals applied to at least one of the first and second combiner inputs,  142  and  143  respectively, and combine signals applied to these first and second combiner inputs at the transmission signal output  53  at levels within the third amplitude range such that the first and second transmission leakage signals arrive at the transmission signal output at levels within an amplitude tolerance range and the total relative phase shift within a phase tolerance range situated about 180 degrees to effect the destructive combination resulting in at least partial cancellation of the first and second transmission leakage signals at the transmission signal output; and the third and fourth divided transmission signals, when present, arrive at the transmission signal output  53  to effect the constructive combination of the third and fourth divided transmission signals to output the portion of the second transmission signal at the transmission signal output  53 . 
     The signal routing assembly  50  of  FIG. 17  is also configured such that the signal divider/combiner  111  produces the third transmission leakage signal, at the first divider/combiner port  131 , which is the portion of the second divided transmission output signal and produces the fourth transmission leakage signal at the second divider/combiner port  133  which is the portion of the first divided transmission output signal. The first routing device  100  receives the third transmission leakage signal at the second port  125  and produces at the third port  127  the third transmission leakage output signal resulting from the third transmission leakage signal and the second routing device  101  receiving the fourth transmission leakage signal at the second port  126  and producing at the third port  128  the fourth transmission leakage output signal resulting from the fourth transmission leakage signal. 
     The signal routing assembly  50  is configured with at least one magnetic biasing devices  400  or  401 , applied to a respective one of the first routing device  100  or the second routing device  101  thereby creating an imbalance in levels between the first divided transmission output signal and the second divided transmission output signal, and an imbalance in levels between the first transmission leakage signal and the second transmission leakage signal. Once the imbalance is created, the signal combiner  112  having the first and second combiner inputs,  142  and  143  respectively, receives the third and fourth transmission leakage output signals, and receives, via of the first and second routing devices,  100  and  101  respectively, the first transmission leakage signal and the second transmission leakage signal. The signal combiner  112  is so configured as to destructively combine at the transmission signal output  53  the third and fourth transmission leakage output signals and to destructively combine the first and second transmission leakage signals to effect at least partial cancellation of the third and fourth transmission leakage output signals and the first and second transmission leakage signals at the transmission signal output. 
     The magnetic biasing device,  400 , or magnetic biasing device,  401 , will be set to effect leakage cancellation such that at least 25 dB isolation is achieved over the desired frequency range. More preferably, a magnetic biasing device should be set to effect leakage cancellation such that at least 30 dB isolation is achieved over the desired frequency range. Still more preferably, a magnetic biasing device should be set to effect leakage cancellation such that at least 35 dB isolation is achieved over the desired frequency range. Still more preferably, a magnetic biasing device should be set to effect leakage cancellation such that at least 40 dB isolation is achieved over the desired frequency range. 
     Circulators as described in this invention are often constructed with ferromagnetic components that may create intermodulation distortion (IMD) of the applied input signal. The signal routing assembly  50  described in this invention is capable of improving the IMD performance over that of a single circulator. It is known by one skilled in the art that the IMD performance is a function of the applied power level. For example, if the power level to a nonlinear device, such as a circulator, is doubled, then the third order IMD product would increase by a factor of four. The same is true for a reduction in the applied power, in that if the power is halved, the third order IMD products would reduce by a factor of four. As the input transmission signal from a transmitter is divided into the first and second divided transmission signals by the signal divider  110 , the power level to each circulator is reduced in half resulting in an improvement of the IMD performance when using the signal routing assembly,  50 . In addition, the balanced structure of the signal routing device  50  will further reduce the level of intermodulation distortion. 
     Another embodiment of the present invention relating to a signal routing assembly makes use of power dividers in place of first and second routing devices to route the signals to and from the common port through the routing device  50 .  FIG. 18  shows the routing device  50  implemented with power dividers  310  and  320 .  FIG. 18  shows power dividers of type Wilkinson tee but it is known that other power dividers can be used such as quadrature hybrids, branch-line couplers, Lange couplers, resistive dividers, T-junctions, reactive T, rat-race, ring hybrids, and other type of hybrid dividers or power dividers that split the power by a pre-determined ratio. The mathematical analysis using power dividers in place of circulators is similar to the derivation for the configuration shown in  FIG. 6  and  FIG. 7 . One of the key differences when using power dividers such as a Wilkinson tee divider in place of circulators is an additional reduction in the amplitude of the signals as they pass from connecting lines  121  and  122  to connecting lines  130  and  132  respectively as shown on  FIG. 18 . As an example using an ideal equal-split Wilkinson tee, a signal passing through power divider  310  from connecting line  121  to connecting line  130  will experience a 3 dB loss in power as half of the power is absorbed in resistor  314 . A signal passing through power divider  320  from connecting line  122  to connecting line  132  will experience a 3 dB loss in power as half of the power is absorbed in resistor  324 . The ideal equal-split power divider assumes no insertion loss in the path. In practice, the insertion loss would add to the 3 dB power divider loss. Typical insertion loss values could range from 0.1 dB to 3 dB depending on the construction and material used in the power divider. In addition, there is an additional reduction in the amplitude of the signals as signals pass from connecting lines  130  and  132  to connecting lines  140  and  141  respectively as shown on  FIG. 18 . As an example using an ideal equal-split Wilkinson tee, a signal passing through power divider  310  from connecting line  130  to connecting line  140  will experience a 3 dB loss in power. A signal passing through power divider  320  from connecting line  132  to connecting line  141  will experience a 3 dB loss in power. 
     Power dividers may also be designed with unequal power splits. For example, the routing device  50  in  FIG. 18  a signal entering port  313  of power divider  310  can be configured so the power leaving port  311  will be larger than the power leaving port  312 . The power divider  310  can also be configured so the power leaving port  311  will be smaller than the power leaving port  312 . As the routing device  50  is generally configured in a symmetrical manner, it is expected that power divider  320  would be configured in the same configuration as power divider  310 . For example, if power divider  310  was configured with power leaving port  311  to be larger than  312  with a pre-defined ratio then power divider  320  would be configured to have the power leaving port  321  to be larger by port  322  by the same ratio. 
     Also note that practical power dividers have undesired leakage paths between the ports  311  to port  312  and port  321  to port  322 . As in the case when routing devices are circulators, the routing device  50  of  FIG. 18  is capable of canceling the undesired leakage from port  51  to port  144  and allowing the leakage energy to be absorbed in the termination  146 . Routing device  50  is also capable of canceling the undesired leakage between port  144  to port  51  and allowing the leakage energy to be absorbed in the termination  120 . 
     The electromagnetic signal routing assembly  50  for effecting two way duplex transmissions of  FIG. 18  can be further described as comprising of the transmission signal input  51  for receiving first transmission signal, the common port  52  for outputting the portion of the first transmission signal from the transmission signal input  51  and receiving, when present, the second transmission signal, and the transmission signal output  53  for outputting the portion of the second transmission signal when present. The signal divider  110  receives the first transmission signal from the transmission signal input  51  and divides the first transmission signal into the first and second divided transmission signals output from ports  118  and  117  respectively and having amplitudes within a first amplitude range of each other and a first relative phase shift therebetween. 
     The first routing device, power divider  310 , has first, second and third ports, namely  311 ,  313  and  312  respectively. The power divider  310  is configured to deliver a portion of signal at the first port  311  to the second port  313  and also is configured to deliver another signal, when present, at the second port  313  to the third port  312 . 
     The second routing device, power divider  320 , has first, second and third ports, namely  321 ,  323  and  322  respectively. The power divider  320  being configured to deliver a portion of signal at the first port  321  to the second port  323  and also being configured to deliver another signal, when present, at the second port  323  to the third port  322 . 
     For both power dividers,  310  and  320 , the following transmission coefficients are defined as
         s 21  is a transmission coefficient from the first port to the second port;   s 32  is a transmission coefficient from the second port to the third;   s 31  is a transmission coefficient from the first port to the third port;   s 21  is greater than s 31 ; and   s 32  is greater than s 31 ;       

     The first power divider  310  has first divided transmission signal applied to the first port  311  of the power divider  310  and simultaneously produces the first divided transmission output signal at the second port  313  of the power divider  310  and the first transmission leakage signal at the third port  312  of the power divider where each are resultant from the first divided transmission signal. 
     The power divider  310  also has the third divided transmission signal, when present, applied to the second port  313  of the power divider  310  and produces, simultaneously with the first divided transmission output signal and the first transmission leakage signal, the third divided transmission output signal at the third port  312  of the power divider  310  resultant from the third divided transmission signal. The power divider  320  has the second divided transmission signal applied to the first port  321  of the power divider  320  and simultaneously produces the second divided transmission output signal at the second port  323  of the power divider  320  and the second transmission leakage signal at the third port  322  of the power divider  320 , each resultant from the second divided transmission signal. The power divider  320  also has the fourth divided transmission signal, when present, applied to the second port  323  of the power divider  320  and produces, simultaneously with the second divided transmission output signal and the second transmission leakage signal, the fourth divided transmission output signal at the third port  322  of the power divider  320  resultant from the fourth divided transmission signal. 
     The signal divider/combiner  111  has the first and second divider/combiner ports  131  and  133  respectively for receiving the first and second divided transmission output signals and the configuration for combining these signals at levels within the second relative amplitude range and with the second relative phase shift therebetween to output the portion of the first transmission signal to the common port  52  and the configuration being such that the second transmission signal, when present and received at the common port  52 , the second transmission signal is divided into the third and fourth divided transmission signals having amplitudes within the second relative amplitude range and with the second relative phase shift therebetween. 
     The signal combiner  112  has first and second combiner inputs,  142  and  143  respectively, for receiving the first and second transmission leakage signals and, when present, for receiving the third and fourth divided transmission output signals. The signal combiner  112  includes the transmission signal output  53  and the signal combiner  112  is configured to introduce the third relative phase shift into signals applied to at least one of the first and second combiner inputs  142  and  143 , and combine signals applied to the first and second combiner inputs,  142  and  143 , at the transmission signal output at levels within the third amplitude range such that the first and second transmission leakage signals arrive at the transmission signal output  53 , at levels within an amplitude tolerance range and a total relative phase shift therebetween within a phase tolerance range situated about 180 degrees, to effect a destructive combination resulting in at least partial cancellation of the first and second transmission leakage signals at the transmission signal output  53  and the third and fourth divided transmission signals, when present, arrive at the transmission signal output  53  to effect a constructive combination of the third and fourth divided transmission signals to output the portion of the second transmission signal at the transmission signal output  53 . 
     Referring to  FIG. 19 , another aspect of the present invention provides that routing device  50  may also be operated as a power divider with high isolation or a power combiner with high isolation. For example, when routing device  50  is operated as a power divider with high isolation and using equal-split power dividers for power dividers  310  and  320 , a signal entering the common port  52  of routing device  50  will split equally between the transmission signal input  51  and transmission signal output  53  which function as first and second output ports. The generalized construction of the equal-split power divider is shown in  FIG. 19 . The routing device  50  configured as an equal-split power divider has one input port at the common port  52 , a first divider signal output port at the transmission signal input  51  and a second divider output signal port at the transmission signal output  53 . For purposes of clarity these alternative names for inputs and outputs will be used in the following discussion in association with reference designators  50 ,  51  and  52 . Similarly, port names and component names are modified and introduced in association with prior reference designators hereinafter to better correspond to functioning of the routing assembly  50  as a power divider. The input signal is received at the input port  52  and routed to the input  134  of the signal divider  111 . The input port  52  is typically connected to a local transmitter or signal source. Signal divider  111  divides the transmission signal into first and second divided signals output at ports  131  and  133  and having substantially equal amplitudes and a first relative phase shift therebetween. The signal divider  111  is any of a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one output, or other device so functioning to divide a signal. The generalized construction of the routing device  50  shown in  FIG. 19  may also be operated as a high isolation power combiner. Two signals entering port  51  and  53  respectively will be combined and output at port  52 . Leakage of power divider  310  between ports  311  and  312  and leakage of power divider  320  between ports  321  and  322  will be canceled. 
     The first and second divided signals are routed to first and second power dividers,  310  and  320 , each having at least first, second and third ports. The divided signals enter the first ports and are routed to the second and third ports, the outputs of which are applied to the signal combiners  110  and  112 . The signal combiners  110  and  112  are any of a quadrature hybrid, or an equal phase power splitter, e.g., a Wilkinson power splitter, a resistive divider, a T-junction or a reactive T, with a phase shift network applied to one output, or other device so functioning to combine the signals from the second and third ports of power dividers  310  and  320 . The combined signals from signal combiner  110  exit the signal combiner  110  at port  116  as a combined signal and is routed to the first divided signal output port  51 . The combined signal from signal combiner  112  exits the signal combiner  112  at port  144  and is routed to the second divided signal output port  53 . 
       FIG. 19  shows routing device  50  for routing signals that travel from input port  52  to the first and second divider signal output ports  51  and  53 . The complex input signal S 60  will be assumed to have voltage amplitude equal to 1 and phase equal to 0 degrees. TABLE 6 summarizes the amplitudes and relative phases for the signals traveling from the input port  52  to the first and second divider signal output ports  51  and  53 . As shown in  FIG. 19 , the input signal S 60  enters the quadrature hybrid  111  at port  134  and the signal is divided into two equal amplitude signals with quadrature phase. The signal S 62  leaving port  133  has amplitude equal to 1/sqrt(2) and relative phase equal to −90 degrees and the signal S 61  leaving port  131  has amplitude equal to 1/sqrt(2) and relative phase equal to 0 degrees. The quadrature hybrid  111  can also be configured with these two connections swapped. In this case, the connections to the other two quadrature hybrids would also need to be swapped in order to maintain the same performance. The two output signals from the quadrature hybrid  111  travel along connecting lines  132  and  130  respectively. The length of transmission line for connecting lines  132  and  130  introduce an additional phase shift of −φ1 to each signal S 62  and S 61 . Signal S 61  enters power divider  310  and is divided in half into S 63  and S 64 . The amplitude of signals S 63  and S 64  is ½. Power divider  310  introduces an additional phase shift of −φ2 into S 63  and S 64 . Signal S 62  enters power divider  320  and is divided in half into S 65  and S 66 . The amplitude of signals S 65  and S 66  is ½. Power divider  320  introduces an additional phase shift of −φ2 into S 65  and S 66 . Connecting lines  121  and  122  introduce and additional phase shift −φ3 to signals S 63  and S 66 . Connecting lines  140  and  141  introduce and additional phase shift −φ4 to signals S 64  and S 65 . Signals S 63  and S 66  enter quadrature hybrid  110  at port  118  and  117  respectively. Each input signal to the quadrature hybrid  110  is divided in half. A relative phase shift of −90 degrees is introduced into the signal passing from the port  118  over to the port  116 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  117  over to the port  119 . Input signals S 63  and S 66  constructively add to output a signal S 67  at port  116  of quadrature hybrid  110 . Signals S 64  and S 65  enter quadrature hybrid  112  at port  142  and  143  respectively. Each input signal to the quadrature hybrid  112  is divided in half. A relative phase shift of −90 degrees is introduced into the signal passing from the port  142  over to the port  144 . A relative phase shift of −90 degrees is introduced into the signal passing from the port  143  over to the port  145 . Input signals S 64  and S 65  constructively add to output a signal S 68  at port  144  of quadrature hybrid  112 . Table 6 shows the amplitude and relative phase relationship between signals shown in  FIG. 19 . While not a requirement the phase shift, −φ3, introduced in connecting lines  121  and  122  would be approximately the equal to the phase shift, −φ4, introduced in connecting lines  140  and  141 . Also, while not a requirement, the phase shift introduced by connecting line  115  would be the same as the phase shift introduced by connecting line  147 . Having all the phase shifts of the respective connecting lines would allow the output signals from port  51  and  53  to have approximately equal amplitude and equal relative phase shift. It is possible to configure the routing device  50  to have an unequal power split between the output signal  51  and  53  relative to the input signal  52 . In this case, power dividers  310  and  320  will be designed to have an unequal power split. For example, if a +6 dB relative power ratio between the signals from port  51  and  53  is required then the relative power ratio between ports  311  and  312  of divider  310  would be +6 dB and the relative power ratio between port  321  and port  322  of divider  320  would also be +6 dB. In practice, large values in power ratio are difficult to achieve as the required transmission line impedance are too high to practically implement. Power ratio ranges between 0 dB (equal power split) to approximately 6 dB are possible. 
     
       
         
           
               
               
               
               
             
               
                   
                 TABLE 6 
               
               
                   
                   
               
               
                   
                 Signal 
                 Amplitude 
                 Phase 
               
               
                   
                   
               
             
            
               
                   
                 S60 
                 1 
                  0 
               
               
                   
                 S61 
                 1/sqrt(2) 
                  0 
               
               
                   
                 S62 
                 1/sqrt(2) 
                 −90 
               
               
                   
                 S63 
                 1/2 
                 −φ1 − φ2 − φ3 
               
               
                   
                 S64 
                 1/2 
                 −φ1 − φ2 − φ4 
               
               
                   
                 S65 
                 1/2 
                 −90 − φ1 − φ2 − φ4 
               
               
                   
                 S66 
                 1/2 
                 −90 − φ1 − φ2 − φ3 
               
               
                   
                 S67 
                 1/sqrt(2) 
                 −90 − φ1 − φ2 − φ3 
               
               
                   
                 S68 
                 1/sqrt(2) 
                 −90 − φ1 − φ2 − φ4 
               
               
                   
                   
               
            
           
         
       
     
     It will also be appreciated in view of this disclosure that practical limitations in the performance of the divider/combiner  111  may introduce undesired leakage signals that may reduce the isolation between port  51  to port  53  and port  53  to port  51  of routing device  50  when first and second power dividers,  310  and  320 , are used. This effect was previously examined with improvements in isolation being achieved with a reflector device placed along connecting line  137  or connecting line  130  or connecting line  132  in  FIG. 18 . For a reflector device placed along line  130 , the power division of the power divider  310  would reduce the amplitude level of the divider/combiner leakage signal, Ls, by a factor of K. The associated compensating signal, Cs, would require a reduction in amplitude by the same amount of K. For example, when using equal-split power dividers, the factor of K would equal 1/sqrt(2) which is equivalent to a 3 dB power division ratio. In this case the amplitude of the leakage signal, Ls, would be HK. The associated compensating signal, Cs, would also require a reduction in amplitude by the same amount of K in this case, XK/2. As both the leakage signal, Ls, and compensating signal, Cs, experience the same reduction in amplitude, the results would be the same as found in the routing device  50  using circulators. The same would be required for a reflector device placed along line  132 . The associated compensating signal, Cs, would also require a reduction in amplitude by the same amount of K. In this case the amplitude of the leakage signal, Ls, would be HK. The associated compensating signal, Cs, would also experience the same reduction in amplitude of factor K resulting is a signal amplitude of YK/2. As both the leakage signal, Ls, and compensating signal, Cs, experience the same reduction in amplitude, the results would be the same as found in the routing device  50  using circulators. For a reflector device placed along connecting line  137 , both the leakage signal, Ls, and compensating signal, Cs, have the same reduction in amplitude by factor K, the results would be the same as found in the routing device  50  using circulators. A reflector device will be set to effect cancellation of the divider/combiner leakage signal, Ls, such that the isolation between port  51  to port  53  is at least 25 dB is achieved over a frequency range associated with the system use. More preferably, a reflector device should be set to effect leakage cancellation such that at least 30 dB isolation is achieved over the desired frequency range. Still more preferably, a reflector device should be set to effect leakage cancellation such that at least 35 dB isolation is achieved over the desired frequency range. Still more preferably, a reflector device should be set to effect leakage cancellation such that at least 40 dB isolation is achieved over the desired frequency range. 
     It will also be appreciated in view of this disclosure that practical limitations in the performance of the divider/combiner  111  may introduce undesired leakage signals that may reduce the isolation between port  51  to port  53  and port  53  to port  51  of routing device  50  when first and second routing devices,  155  and  156 , are directional couplers as shown in  FIG. 10 . Improvements in isolation can be achieved with a reflector device placed along connecting line  130  or connecting line  132  or connecting line  137 . A reflector device will be set to effect cancellation of the divider/combiner leakage signal, Ls, such that the isolation between port  51  to port  53  is at least 25 dB is achieved over a frequency range associated with the system use. More preferably, a reflector device should be set to effect leakage cancellation such that at least 30 dB isolation is achieved over the desired frequency range. Still more preferably, a reflector device should be set to effect leakage cancellation such that at least 35 dB isolation is achieved over the desired frequency range. Still more preferably, a reflector device should be set to effect leakage cancellation such that at least 40 dB isolation is achieved over the desired frequency range. 
     Having described preferred embodiments of the invention with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments, and that various changes and modifications may be effected therein by one skilled in the art without departing from the scope or spirit of the invention as defined in the appended claims. Such modifications include substitution of components for components specifically identified herein, wherein the substitute component provide functional results which permit the overall functional operation of the present invention to be maintained. Such substitutions are intended to encompass presently known components and components yet to be developed which are accepted as replacements for components identified herein and which produce result compatible with operation of the present invention. Furthermore, while examples have been provided illustrating operation at certain power levels and frequencies, the present invention as defined in this disclosure and claims appended hereto is not considered limited to frequencies and power levels recited herein. It is furthermore to be understood that the receiver and transmitter referenced herein is not considered limited to any particular types of receivers or transmitters nor any particular form of signals in that the signals may carry analog or digital information, in any modulation scheme, or the signals need not carry information. Furthermore, the signals used in this invention are considered to encompass any electromagnetic wave transmission.