Patent Publication Number: US-7915879-B2

Title: Switching converter including a rectifier element with nonlinear capacitance

Description:
BACKGROUND 
     A switching converter serves for converting an input voltage, which can be applied to input terminals, into an output voltage, which is available at output terminals. There are various switching converter topologies. What is common to these switching converter topologies is that at least the following circuit components are present: an inductive storage element, a switching element, a rectifier element and a capacitive storage element. During the operation of the switching converter, the switching element is switched on and off cyclically. The individual circuit components are connected up in such a way that the inductive storage element buffer stores energy during switched-on phases of the switching element and outputs at least part of the stored energy during a subsequent switched-off phase to the capacitive storage element via the rectifier element. 
     Transitions from a switched-on phase to a switched-off phase of the switching element are critical with regard to electromagnetic interference radiation and critical with regard to overvoltages that can occur at parasitic inductances if a current flowing through the switching element changes rapidly. During such a transition phase, a current through the inductive storage element that flows through the switching element during the switched-on phase has to be accepted by the rectifier element. Extremely high changes in voltages present across the switching element and the rectifier element, and in currents flowing through these components can occur during this transition. 
     SUMMARY 
     One embodiment of the present description provides a switching converter including a switching element, which can be driven in the on state and in the off state. A first capacitive element is between the load path terminals of the switching element and has a first capacitance having a nonlinear capacitance characteristic curve that is dependent on a voltage between the load path connections. A rectifier element connected between the inductive storage element and the capacitive storage element configure such that it enables a current flow between the inductive storage element and the capacitive storage element when the switching element is driven in the off state. A second capacitive element is between the load path terminals of the rectifier element and has a second capacitance having a nonlinear capacitance characteristic curve that is dependent on a voltage between the load path connections. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings are included to provide a further understanding of embodiments and are incorporated in and constitute a part of this specification. The drawings illustrate embodiments and together with the description serve to explain principles of embodiments. Other embodiments and many of the intended advantages of embodiments will be readily appreciated as they become better understood by reference to the following detailed description. The elements of the drawings are not necessarily to scale relative to each other. Like reference numerals designate corresponding similar parts. 
         FIG. 1  illustrates one embodiment of a switching converter on the basis of an electrical equivalent circuit diagram. 
         FIG. 2  illustrates temporal profiles of some signals that occur in the switching converter. 
         FIG. 3  illustrates the temporal profile of a voltage present across a switching element of the switching converter in the case of the transition from a switched-on phase to a switched-off phase of the switching element. 
         FIG. 4  illustrates capacitance characteristic curves of capacitive components present in the switching converter. 
         FIG. 5  illustrates one embodiment of a MOS transistor that functions according to the compensation principle, on the basis of a cross section through a semiconductor body. 
         FIG. 6  illustrates one embodiment of a bipolar diode that functions according to the compensation principle, on the basis of a cross section through a semiconductor body. 
         FIG. 7  illustrates one embodiment of a Schottky diode that functions according to the compensation principle, on the basis of a cross section through a semiconductor body. 
         FIG. 8  partially illustrates an electrical equivalent circuit diagram of a switching converter modified by comparison with the switching converter in accordance with  FIG. 1 . 
         FIG. 9  illustrates one embodiment of a switching converter on the basis of an electrical equivalent circuit diagram. 
         FIG. 10  illustrates one embodiment of a switching converter on the basis of an electrical equivalent circuit diagram. 
         FIG. 11  illustrates one embodiment of a switching converter on the basis of an electrical equivalent circuit diagram. 
         FIG. 12  illustrates the voltage-dependent profile of an effective capacitance present at a circuit node in the switching converter. 
     
    
    
     DETAILED DESCRIPTION 
     In the following Detailed Description, reference is made to the accompanying drawings, which form a part hereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. In this regard, directional terminology, such as “top,” “bottom,” “front,” “back,” “leading,” “trailing,” etc., is used with reference to the orientation of the Figure(s) being described. Because components of embodiments can be positioned in a number of different orientations, the directional terminology is used for purposes of illustration and is in no way limiting. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. The following detailed description, therefore, is not to be taken in a limiting sense, and the scope of the present invention is defined by the appended claims. 
     It is to be understood that the features of the various exemplary embodiments described herein may be combined with each other, unless specifically noted otherwise. 
       FIG. 1  illustrates one embodiment of the electrical equivalent circuit diagram of a switching converter, which is embodied as a boost converter in the example illustrated. This switching converter has input terminals  11 ,  12  for applying an input voltage Vin and output terminals  13 ,  14  for proving an output voltage Vout. In the switching converter illustrated by way of example, one  12  of the input terminals and one  14  of the output terminals are at a common electrical potential, for example a reference potential, such as e.g., ground. 
     The switching converter has an inductive storage element  21 , a switching element  22 , a rectifier element  24  and a capacitive storage element  26 . The inductive storage element  21  can be in one embodiment a choking coil having an inductor core (not illustrated). The capacitive storage element  26  is connected between the output terminals  13 ,  14  and serves for providing the output voltage Vout. This capacitive storage element  26  is realized as a capacitor, for example, and is also referred to hereinafter as output capacitor of the switching converter. The switching element  22  has a control terminal  221  for feeding in a control signal S 22  and also load path terminals  222 ,  223  and a load path running between the load path terminals  222 ,  223 . The switching element  22  can be driven in the on state and in the off state, or switched on and off, by using the control signal S 22  and is connected up in such a way that the inductive storage element  21  can take up electrical energy via the input terminals  11 ,  12  when the switching element  22  is switched on. In the boost converter illustrated in  FIG. 1 , the load path of the switching element  22  is connected in series with the inductive storage element  21  for this purpose, wherein the series circuit including the inductive storage element  21  and the switching element  22  is connected between the input terminals  11 ,  12 . 
     The rectifier element  24  is connected up in such a way that, when the switching element  22  is driven in the off state, the rectifier element enables a current flow from the inductive storage element  21  to the capacitive storage element  26 , but prevents a current flow from the output capacitor in a direction of the inductive storage element. In the example illustrated, for this purpose the rectifier element  24  is connected between the inductive storage element  21  and the capacitive storage element  26 . In the example, the rectifier element  24  and the capacitive storage element  26  form a series circuit connected in parallel with the switching element  22 . 
     The rectifier element  24  has load path terminals  241 ,  242  with a load path running between the load path connections  241 ,  242 . The rectifier element is a diode, for example, which is connected in the forward direction between the inductive storage element  21  and the capacitive storage element  26 . In this case, a cathode terminal of the diode forms the first load path terminal  241 , and an anode terminal of the diode forms the second load path terminal  242 . Even though the circuit symbol of a bipolar diode is illustrated in the electrical equivalent circuit diagram in  FIG. 1 , it should be pointed out that either a bipolar diode or a Schottky diode can be used as the diode. 
     The switching element  22  can be driven in a conventional manner by a drive circuit  100  (illustrated by dashed lines), which provides a pulse-width-modulated signal that is fed to the control terminal  221  of the switching element  22  as switching signal S 22 . 
     The basic functioning of the switching converter illustrated in  FIG. 1  is explained below with reference to  FIG. 2 , which illustrates by way of example temporal profiles of the following signals: of a current I 21  through the inductive storage element  21 , which is also referred to hereinafter as inductance current; of the drive signal S 22 ; of a voltage V 22  across the load path of the switching element  22 , which is also referred to hereinafter as first load path voltage; and a voltage V 24  across a load path of the rectifier element  24 , which is also referred to hereinafter as second load path voltage. It shall be assumed for the explanation below that the voltage V 24  across the rectifier element is a positive voltage if the rectifier element is reverse-biased when the switching element  22  is driven in the on state. The rectifier element illustrated is reverse-biased if a positive voltage is present between the first and second load path connections. 
     For further explanation, a drive period of the switching element  22  shall be considered, during which drive period the switching element  22  is driven in the on state or switched on, for a switched-on duration Ton, and is driven in the off state, or switched off, for a switched-off duration Toff. The time duration during which the switching element  22  is switched on is also referred to hereinafter as switched-on phase; the time duration during which the switching element  22  is switched off is also referred to hereinafter as switched-off phase. 
     If the ohmic resistance of the switching element  22  in the switched-on state is disregarded, then approximately the entire input voltage Vin is present across the inductive storage element  22  during the switched-on phase. A current I 21  through the inductive storage element  21  arises during this switched-on phase. The inductive storage element  21  is magnetized during this switched-on phase, that is to say that electrical energy is stored in the inductive storage element  21 . A change in the current I 21  with respect to time is dependent on the input voltage during this switched-on phase and is all the greater, the greater the amplitude of the input voltage Vin is. As long as a magnetization state of the inductive storage element  21  remains below a saturation region, the change in the current with respect to time is approximately proportional to the input voltage, which is taken as a basis for the illustration in accordance with  FIG. 1 . The rectifier element  24  is reverse-biased during the switched-on phase. If the ohmic resistance of the switching element  22  during the switched-on phase is disregarded, then approximately the output voltage Vout is present across the rectifier element  24  during the switched-on phase. 
     If the switching element  22  is driven in the off state, then the electrical potential at the node common to the inductive storage element  21  and the rectifier element  24  rises until the rectifier element  24  is forward-biased and thus enables a current flow from the inductive storage element  21  to the capacitive storage element  26 . The electrical potential at the node common to the inductive storage element  21  and the rectifier element  24  then corresponds to the sum of the forward voltage of the rectifier element  24  and the output voltage Vout. In bipolar diodes or Schottky diodes, the forward voltage is at most in the region of a few volts and is usually negligibly small in comparison with the output voltage Vout, which can be in the region of a few hundred volts. A boost converter illustrated in  FIG. 1  serves for example for converting an input voltage Vin resulting from a mains voltage into a intermediate circuit voltage. In one embodiment, values for the intermediate circuit voltage are in this case in the region of 400V. 
     The current I 21  through the inductive storage element  21  decreases during the switched-off phase. In this case, a change in the current I 21  with respect to time during the switched-off phase is dependent on a difference between the input voltage Vin and the output voltage Vout. Overall, the current I 21  through the inductive storage element  21  has a triangular current profile. In this case, the switching element  22  can be driven in such a way that the current decreases to zero between two drive periods. In this context this is referred to as a discontinuous current mode (DCM). However the switching element  22  can also be driven in such a way that the current I 21  is always greater than zero. In this context this is referred to as a continuous current mode (CCM). 
     The drive circuit  100  can vary the duty cycle of the drive signal S 22  and/or the switching frequency with which the switching element is switched on and off, in order to regulate the output voltage Vout largely independently of the current taken up by a load Z that can be connected to the output terminals  13 ,  14  (illustrated by dashed lines). For this purpose, information about the output voltage Vout is fed to the drive circuit  100 . A decrease in the output voltage Vout as the current taken up by the load Z increases can be counteracted, for example, by increasing the duty cycle, that is to say increasing the switched-on duration Ton in comparison with the total duration Tp of the drive period. An increase in the output voltage Vout as the current taken up by the load Z decreases can be correspondingly counteracted by reducing the duty cycle, that is to say by reducing the switched-on duration Ton in comparison with the total duration Tp. Such regulating methods either for a DC mode or a CC mode are sufficiently known, such that further more detailed explanations in this respect are not required. 
     Irrespective of the operating mode (CCM or DCM) and irrespective of specific regulating methods for regulating the output voltage Vout, the transition of the switching element  22  from the switched-on phase to the switched-off phase can be critical with regard to overvoltage spikes. During the transition from the switched-on phase to the switched-off phase, referring to  FIG. 2 , the load path voltage V 22  of the switching element  22  arises from a minimum value V 22   min  to a maximum value V 22   max . In this case, the minimum value V 22   min  corresponds to the voltage drop across the switching element  22  driven in the on state. The switching element  22  is for example a power MOS transistor having a dielectric strength of a few hundred volts, for example 600V. The voltage drop of such a component is in the region of a few volts if the component is driven in the on state. The maximum value V 22   max  of the load path voltage V 22  of the switching element  22  is in the region of the output voltage Vout, which is for example in the region of few hundred volts. 
     I 21   max  shall hereinafter designate the value of the current through the inductive component  21  which is reached at the instant when the switching element  22  is switched off. The inductance current I 21  flows via the switching element during the switched-on phase, such that during the time during which the load path voltage V 22  of the switching element  22  rises from the minimum value V 22   min  to the maximum value V 22   max , a current I 22  through the switching element  22  decreases from the maximum value I 21   max  to zero. During the switched-off phase, the inductance current I 21  flows via the rectifier element  24 , where the rectifier element  24  can accept the maximum current I 21   max  after the changeover instant only when the first load path voltage V 22  has risen up to a value corresponding to the sum of a forward voltage of the rectifier element  24  and the output voltage Vout. If the first load path voltage V 22  rises very rapidly, then a “hard” change occurs in the inductance current I 21  from the current path in which the switching element  22  is arranged into the current path in which the rectifier element  24  is arranged. Such a hard change, that is to say a large change in the current flowing through the rectifier element  24  with respect to time, can lead to voltage spikes at parasitic inductances present in the current path with the rectifier element  24 . The following applies for such voltage spikes: 
     
       
         
           
             
               
                 
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     In this case, V denotes the amplitude of the voltage spike, L denotes the inductance value of the parasitic inductances and dI 24 /dt denotes the change in the current I 24  flowing through the rectifier element  24  with respect to time. Parasitic inductances are not explicitly illustrated in  FIG. 1 . Such parasitic inductances can already be formed by line connections in the current path in which the rectifier element  24  is present. 
     It follows from the equation (1) that the amplitude of voltage spikes that occur is directly dependent on the change in the current I 24  with respect to time. In order to reduce such voltage spikes it is desirable to “flatten” the voltage rise in the first load path voltage V 22  toward the end of the transition phase, that is to say shortly before the rectifier element  24  starts to conduct. Furthermore, it is desirable to “flatten” the voltage rise in the load path voltage V 22  at the beginning of the transition phase as well. 
     It is therefore endeavored to cause the rise in the first load path voltage V 22  during the transition phase to proceed in three phases: a first phase, in which the voltage V 22  rises from the minimum value V 22   min  up to a first voltage value V 22   1 ; a second phase, in which the load path voltage V 22  rises from the first value V 22   1  to a second value V 22   2 ; and a third phase, in which the load path voltage V 22  rises from the second value V 22   2  to the maximum value V 22   max . Such a rise in the first load path voltage V 22  in three phases is illustrated in  FIG. 3 . A rate of rise, that is to say a change dV 22 /dt in the load path voltage V 22  with respect to time, is in this case intended to proceed more slowly during the first and third phases than during the second phase. In a switching converter having an output voltage Vout of 400V, for example, the first voltage value V 22   1 , which marks the end of the first phase is 100V, for example, and the second value V 22   2  which marks the end of the second phase, is 300V, for example. Generally, the first voltage value V 22   1  is for example between 10% and 30% of the maximum value V 22   max  and the second voltage value V 22   2  is for example between 70% and 90% of the maximum value V 22   max . It should be pointed out that the rate of voltage rise does not have to be constant within the individual phases. The rates of voltage rise that occur during the first and third phases are lower, however, than the rates of voltage rise that occur during the second phase. 
     Such a profile of the voltage rise in the first load path voltage V 22  in three phases can be achieved by virtue of the fact that a capacitive element  23  having a nonlinear capacitance characteristic curve that is dependent on the voltage V 22  between the load path connections  222 ,  223  of the switching element  22  is provided between the load path connections  222 ,  223 . The capacitance characteristic curve is determined by a dependence of a capacitance C 23  of the first capacitive element  23  on the first load path voltage V 22 . Referring to  FIG. 4 , the capacitance characteristic curve is for example such that the capacitance C 23 , starting from a voltage value at which the capacitance C 23  has its maximum value C max  and which is zero, for example, decreases for increasing voltage values. 
     A second capacitive element  25  is present between the load path connections  241 ,  242  of the rectifier element  24 , the second capacitive element having a nonlinear capacitance characteristic curve that is dependent on the voltage V 24  between the load path connections  241 ,  242 . The capacitance characteristic curve is determined by a dependence of a capacitance C 25  of the second capacitive element  25  on the second load path voltage V 24 . The profile of the capacitance characteristic curve depending on the second load path voltage V 24  corresponds qualitatively to the profile of the capacitance characteristic curve of the first capacitive element  23  depending on the voltage V 22 , such that reference is made to the explanation given above. It should be noted in this context that maximum capacitance values of the first and second capacitive elements  23 ,  24  and minimum capacitance values of the capacitive elements  23 ,  24  can differ. Furthermore, the limit value of the voltages V 22 , V 24  at which the capacitances C 23 , C 25  respectively assume their minimum values can also differ. Merely for reasons of simplified illustration, identical maximum values C max  and minimum values C min  and identical threshold values V 0  for the capacitances C 23 , C 25  are assumed in  FIG. 4 . The effects of such capacitive elements  23 ,  25  having a voltage-dependent nonlinear capacitance characteristic curve on the voltage rise in the voltage V 22  are explained below. 
     I 21   max  hereafter denotes the value of the inductance current I 21  at the instant when the switching element  22  is switched off. Starting from an instant at which the switching element  22  turns off, a rise in the load path voltage V 22  is then dependent on this maximum current I 21   max  and the capacitance value C 23  of the first capacitive element. The following applies here: 
     
       
         
           
             
               
                 
                   
                     
                       
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     In this case, dV 22 /dt denotes the change in the load path voltage V 22  with respect to time, that is to say the rate at which the load path voltage V 22  rises. In this case, this rate of rise is all the greater, the greater the maximum value I 21   max  of the inductance current I 21  is and the lower the capacitance C 23 , the capacitance being dependent on the first load path voltage is. 
     On account of the explained nonlinear dependence of the capacitance C 23  on the load path voltage V 22  with a large capacitance value for small load path voltages V 22 , the rate of voltage rise dV 22 /dt is lower for small load path voltages V 22  than for larger load path voltages V 22 . If the voltage profile in  FIG. 3  is considered, than the first voltage value V 22   1 , which marks the end of the first phase, corresponds to the threshold value V 0  starting from which the capacitance C 23  assumes it minimum value C min . 
     As the first load path voltage V 22  across the switching element  22  rises, the second load path voltage V 24  across the rectifier element  24  decreases. If the load path voltage V 24  decreases to a value starting from which the capacitance C 25  of the capacitive element  25  rises, then a voltage rise in the first load path voltage V 22  is reduced, to be precise because, as the capacitance C 25  of the second capacitive element  25  increases, a discharging process of the capacitive element  25  slows down, whereby the change in the second load path voltage  24  with respect to time also slows down. It should be noted in this context that the first and second load path voltages V 22 , V 24  are directly related to one another by way of the output voltage Vout, where the following applies:
 
 V 22 +V 24 =V out  (3).
 
     Let V 0  be the threshold value starting from which the capacitance C 25  rises as the second load path voltage V 24  decreases. The following then applies for the load path voltage V 22  starting from which the third phase begins, that is to say starting from which a slowing down of the voltage rise beings:
 
 V 22 2   =V out− V   0   (4).
 
     If the threshold value V 0  differs for the first and second capacitances C 23 , C 25 , the threshold value V 0  for the second capacitance C 25  should be inserted into the equation (4). 
     According to one embodiment, a power MOSFET that functions in accordance with the compensation principle is used as the switching element  22 . One example of such a MOSFET is illustrated on the basis of a cross-sectional illustration in  FIG. 5 . In this case,  FIG. 5  illustrates a cross section through a semiconductor body  100  in which the MOSFET is integrated. The MOSFET includes a drift zone  41  of a first conduction type and one or a plurality of compensation zones  42  of a second conduction type, which is complementary to the first conduction type, the compensation zone(s) being arranged in the drift zone  41 . The MOSFET used as the switching element  22  is an n-channel MOSFET for example. In such a MOSFET, the drift zone  41  is n-doped and the compensation zone  42  is p-doped. The MOSFET additionally has a drain zone  48 , which is adjacent to the drift zone  41 , and which is of the same conduction type as the drift zone  41  but doped more highly. It should be noted in this context that instead of a MOSFET, an IGBT can also be used as the switching element  22 . Such an IGBT differs from a MOSFET essentially by virtue of the fact that the drain zone is doped complementarily to the drift zone. 
     At a side of the drift zone  41  which is remote from the drain zone  48 , a body zone  43  is adjacent to the drift zone  41 , and is doped complementarily to the drift zone  41 . The body zone  43  separates the drift zone  41  from a source zone  44 , which is of the same conduction type as the drift zone  41 . A gate electrode  45  is present for controlling a conducting channel in the body zone  43  between the drift zone  41  and the source zone  44 , which gate electrode is arranged adjacent to the body zone  43  and is dielectrically insulated from the body zone  43 , the source zone  44  and the drift zone  41  by a gate dielectric  46 . Contact is made with the source zone  44  and the body zone  43  jointly by a source electrode  47 , whereby the source zone  44  and the body zone  43  are short-circuited. In this way, a diode  49  is formed between the source electrode  47  and the drain zone  48 . This diode, the circuit symbol of which is illustrated in  FIG. 5  and designated by the reference symbol  49 , is also referred to as a body diode. On account of this body diode, an n-channel MOSFET is able to turn on when a positive voltage is applied between the source electrode  47  and the drain zone  48 , without it being necessary for the MOSFET to be driven in the on state via the gate electrode  45  for this purpose. In the case of a p-channel MOSFET, the forward direction of the body diode correspondingly runs from drain to source. 
     The MOSFET has a drain terminal D connected to the drain zone  48 , a source terminal S connected to the source electrode  47 , and a gate terminal G connected to the gate electrode  45 . These terminals are illustrated merely schematically in  FIG. 5 . When an n-channel MOSFET is used as the switching element, the drain terminal D of the MOSFET forms the first load path terminal  222 , the source terminal S forms the second load path terminal  223  and the gate terminal G forms the control terminal  221 . 
     The functioning of the component illustrated in  FIG. 5  is explained below. It shall be assumed for this explanation that the MOSFET is an n-channel MOSFET. The explanation below also applies correspondingly to a p-channel MOSFET, the component zones of which are doped complementarily in comparison with component zones of an n-channel MOSFET. If the polarity of voltages plays a part in the explanation below, then the polarity of the voltages in the case of a p-channel MOSFET should be correspondingly interchanged with respect to voltages in an n-channel MOSFET. 
     The MOSFET  40  illustrated in  FIG. 5  is turned on if a positive voltage is present between drain D and source S, and if at the gate electrode  45  a drive potential is present that is sufficient to form an inversion channel in the body zone  43  between the source zone  44  and the drift zone  41 . In the case of an n-channel MOSFET, the drive potential is an electrical potential that is positive with respect to source potential. The MOSFET is turned off if a positive voltage is present between drain D and source S, and if a drive potential suitable for forming an inversion channel is not present at the gate electrode  45 . In this case, the pn junction between the drift zone  41  and the body zone  43  is reverse-biased, such that a space charge zone is formed in the drift zone  41  proceeding from the pn junction. In the case of a MOSFET, as is known, the drift zone  41  serves to take up a reverse voltage present between drain and source D, S. pn junctions are additionally formed between the compensation zones  42  and the drift zone  41 . In the example illustrated, the compensation zones  42  are connected to the body zone  43 , such that when a reverse voltage is applied, space charge zones are also formed directly proceeding from the pn junctions between the compensation zones  42  and the drift zone  41 . In a manner not illustrated in more specific detail, the compensations zones  42  can also be arranged in floating fashion at the drift zone  41 . In this case, space charge zones then propagate proceeding from the pn junctions between the compensation zones and the drift zone  41  if a space charge zone propagating in the drift zone  41  encroaches on the respective compensation zone. The compensation zones  42  serve, when the component is driven in the off state, for partly or completely compensating for the dopant charges present in the drift zone  41 . As a result of this, in comparison with components without such compensation zones, a higher doping can be provided in the drift zone  41 , whereby the on resistance of the component can be reduced without, however, reducing the dielectric strength. The total net dopant charge present in the compensations zones  42  ideally corresponds to the total net dopant charge present in the drift zone  41 . 
     In connection with the present description, a compensation component should be understood to mean a component having a drift zone in which one or a plurality of compensation zones  42  present complementarily to the drift zone  41  are provided, and in the case of which the net dopant charge present in the compensation zones amounts to between 80% and 120% of the net dopant charge of the drift zone  41 . 
     The pn junctions between the body zone  43  and the compensation zone  42 , on the one hand, and the drift zone  41 , on the other hand, form a capacitance when the component is driven in the off state, the capacitance also being referred to as junction capacitance or space charge zone capacitance. In comparison with conventional components, compensation components have a large-area pn junction, and hence a pronounced junction capacitance. Owing to this large-area pn junction, such components are also referred to as superjunction components. The junction capacitance is dependent on the voltage present between drain D and source S and decreases as the drain-source voltage increases, that is to say the further the space charge zone of the drift zone  41  propagates when the component is in the off state. This junction capacitance can be used directly as the capacitive element  23 . The function of the switching element  22  and of the capacitance  23  that is effective between the load path connections  222 ,  223  of the switching element  22  and has a voltage-dependent nonlinear capacitance characteristic curve can thus be achieved by using a MOSFET that functions according to the compensation principle as the switching element  22 . In this case, the capacitive element  23  is directly part of the MOSFET. 
     It should be noted that  FIG. 5  merely serves for elucidating the basic principle of a compensation MOSFET. It goes without saying that MOSFETs having a geometry that deviates from the geometry illustrated in  FIG. 5  can be used. Thus, by way of example, instead of the illustrated planar gate electrode  45  arranged above a front side of the semiconductor body  100 , a trench electrode arranged in a trench extending into the semiconductor body  100  proceeding from a front side could also be provided. Such component geometries are sufficiently known, and so further explanations in this respect are not required. Finally, it should also be mentioned that the MOSFET in order to increase the current-carrying capacity, has a multiplicity of transistor cells of identical type, each having a body zone  43  and a source zone  44 . In this case, the individual transistor cells are connected in parallel by their source and body zones  44 ,  43  being short-circuited by the source electrode  47 . In this case, the drift zone  41  and the drain zone  48  are common to all the transistor cells. The individual transistor cells can have any conventional cell geometry, such as a hexagonal geometry, for example. In this case, the body zones  43  have a hexagonal geometry in a sectional plane running perpendicular to the plane of the drawing illustrated in  FIG. 5 . The gate electrode  45  is common to all the transistor cells and has cutouts in the region of the source and body zones  44 ,  43 , in the region of which cutouts the source electrode  47  makes contact with the source and body zones  44 ,  43 . 
     A bipolar diode that functions according to the compensation principle can be used as the rectifier element  24 . In this case, the rectifier element  24  and the second capacitive component  25  are realized by a single component, namely the bipolar diode that functions according to the compensation principle.  FIG. 6  illustrates a schematic illustration of an example of such a bipolar diode  50  that functions according to the compensation principle.  FIG. 6  illustrates a cross section through a semiconductor body  100  in which component zones of the bipolar diode are realized. 
     The diode  50  has two emitter zones  53 ,  54 , which are doped complementarily to one another and between which is arranged a drift zone  51  doped more lightly than the emitter zones  53 ,  54 . Compensation zones  52  doped complementarily to the drift zone  51  are arranged in the drift zone  51 , which compensation zones are either arranged in floating fashion in the drift zone  51  or are connected to the emitter zone that is doped complementarily to the drift zone  51  (as illustrated). In the example illustrated, this is the first emitter zone  53 . That one of the two emitter zones  53 ,  54  which is p-doped forms an anode zone of the diode  50 , and the other one of the two emitter zones, which is n-doped, forms a cathode zone. The forward direction of the diode  50  runs from the anode A to the cathode K, that is to say that the component is driven in the on state when a positive voltage is applied between anode A and cathode K, and is driven in the off state when a positive voltage is applied between cathode K and anode A. When the diode  50  illustrated in  FIG. 6  is used as rectifier element  24  in the switching converter in accordance with  FIG. 1 , the diode is to be connected up in such a way that the cathode connection K forms the first load path connection  241  and the anode connection A forms the second load path connection  242 . 
     In the case of a positive second load path voltage V 24 , this diode is reverse-biased. A space charge zone then propagates proceeding from the pn junctions between the first emitter zone  53  and the compensation zone  52 , on the one hand, and the drift zone  51 , on the other hand. In this case, the capacitive element  25  having a voltage-dependent nonlinear capacitance characteristic curve is formed by the junction capacitance of this pn junction, which decreases as the reverse voltage rises or increases as the reverse voltage decreases. 
     When using a compensation MOSFET as the switching element  22  and a compensation bipolar diode  50  as the rectifier element  24 , it is possible to obtain capacitive elements  23 ,  25  having an approximately identical capacitance characteristic curve by using the dimensions and dopings of the drift zones  41 ,  51  and of the compensation zones  42 ,  52  being chosen to be respectively identical for the production of the compensation MOSFET and for the production of the bipolar diode. However, the capacitive components need not necessarily have identical capacitance characteristic curves.  FIG. 12  schematically illustrates the profile of the capacitance value Cg of an effective capacitance that is effective at the circuit node common to the switching element  22  and the rectifier element  24 , assuming that these two components are compensation components having voltage-dependent output capacitances. 
     The effective capacitance is composed of the capacitance values of the voltage-dependent output capacitances  23 ,  25  of the switching element  22  and of the rectifier element  24  together and is plotted as a function of the voltage V 22  present across the switching element  21  in  FIG. 12 . In the case of small voltage values of the voltage V 22 , the capacitance value of the capacitance is crucially determined by the output capacitance  23  of the first semiconductor switching element  21 , which has its maximum value at a voltage V 21  of zero. Correspondingly, in the case of large voltage values of the voltage V 21 , the capacitance value of the effective capacitance is crucially determined by the output capacitance  24  of the rectifier element  24 . The output capacitance assumes its maximum value if the voltage across the rectifier element  24  is zero, that is to say if the voltage V 22  across the first semiconductor switching element  21  corresponds to the output voltage Vout. A minimum of the capacitance value of the effective capacitance is at a voltage V 21  of between zero and the output voltage Vout. 
     The solid line in  FIG. 12  illustrates an “asymmetrical” case where one of the output capacitances—that of the rectifier element  24  in the example—has a smaller maximum capacitance value than the other output capacitance. The dash-dotted line in  FIG. 12  illustrates a “symmetrical” case where both output capacitances have an identical maximum capacitance value. 
     A Schottky diode can also be used instead of a bipolar diode as rectifier element. Referring to  FIG. 7 , which illustrates a schematic illustration of a cross section through a Schottky diode, such a Schottky diode differs from a bipolar diode by virtue of the fact that a metal layer  55  is provided instead of the emitter zone ( 53  in  FIG. 6 ) doped complementarily to the drift zone  41 , which metal layer forms a Schottky junction with the semiconductor material of the drift zone  51 . A suitable material for this is for example platinum (Pt), tungsten (W) or aluminum (Al). The semiconductor body  100  is composed for example of silicon (Si) or silicon carbide (SiC). A Schottky diode has the advantage over a bipolar diode of a lower forward voltage. 
     Referring to  FIG. 8 , which illustrates an excerpt from the electrical equivalent circuit diagram of a boost converter, instead of a diode as rectifier element  24  it is also possible to provide a compensation MOSFET as rectifier element, which compensation MOSFET is connected up in such a way that its body diode acts as a rectifier element and accepts the inductance current  21  when the switching element  22  is driven in the off state. The gate electrode G of the MOSFET can remain unconnected in this case. Furthermore, there is the possibility of such a MOSFET used as rectifier element  24  being operated as an active rectifier element, that is to say of the MOSFET being driven in the on state during the switched-off phase of the switching element  22  starting from an instant from which it is ensured that the switching element  22  is reliably turned off. The MOSFET used as rectifier element  24  can be still be switched off during the above-explained transition phase during which the first load path voltage V 22  rises to approximately the value of the output voltage Vout and during which the second load path voltage V 24  decreases. During this phase, the body diode accepts an inductance current  21  that is possibly already flowing. The MOSFET used as rectifier element  24  can be driven in the on state for example after the conclusion of the transition phase and is in each case driven in the off state before the switching element  22  is driven in the on state at the beginning of a next driving phase. 
     It goes without saying that the concept explained above, namely of providing capacitive elements having voltage-dependent nonlinear capacitance characteristic curves between load paths of a switching element and of a rectifier element in a switching converter, is not restricted to the boost converter explained above. All the explanations given above with regard to the capacitive elements, and in one embodiment with regard to the possibilities for realizing them, apply to any switching converter topologies, such as, for example, buck converters, buck-boost converters or flyback converters. 
       FIG. 9  illustrates the electrical equivalent circuit diagram of a buck converter. In the case of this buck converter, the switching element  22  is connected in series with the inductive storage element  21  and the capacitive storage element  26  between the input terminals  11 ,  12 . As in the boost converter in accordance with  FIG. 1 , the output voltage is present across the capacitive storage element  26  connected between the output terminals  13 ,  14 . The rectifier element  24  is connected in parallel with a series circuit including the inductive storage element  21  and the capacitive storage element  26 . The signal profiles illustrated in  FIG. 2  for the drive signal S 22 , the first load path voltage V 22  and the second load path voltage V 24  correspondingly apply to the buck converter illustrated in  FIG. 9 . During the switched-on phase, the input voltage Vin is present across the series circuit including the switching element  22 , the inductive storage element  21  and the capacitive storage element  26 . In this case, the voltage present across the inductive storage element  21  approximately corresponds to the difference between the input voltage Vin and the output voltage Vout, where the input voltage Vin—in contrast to a boost converter—is greater than the output voltage Vout. The inductive storage element  21  is magnetized during this switched-on phase. During a subsequent switched-off phase, the rectifier element  26  accepts the inductance current I 21  flowing through the inductive storage element  21 . During the switched-off phase, the load path voltage V 22  assumes its maximum value V 22   max  corresponding to the sum of the input voltage Vin and the forward voltage of the rectifier element  24 . During the preceding switched-on phase, the second load path voltage V 24  assumes its maximum value V 24   max  approximately corresponding to the input voltage Vin. 
     In the transition phase from the switched-on phase to the switched-off phase, the first load path voltage V 22  rises to the abovementioned maximum value V 22   max , while the second load path voltage V 24  decreases proceeding from the abovementioned maximum value V 24   max . During this transition phase, the first and second capacitances  23 ,  25  influence the voltage rise in the first load path voltage V 22  in the manner explained, thereby reducing for example voltage spikes which can occur at parasitic inductances in the current path with the rectifier element  24  when the rectifier element  24  accepts the inductance current I 21 . 
     Referring to  FIG. 10 , the concept explained above can also be applied for example to a buck-boost converter. With regard to its topology, the buck-boost converter differs from the buck converter illustrated in  FIG. 9  by virtue of the fact that the positions of the inductive storage element  21  and of the rectifier element  24  in the circuit topology are interchanged. The inductive storage element  21  is therefore connected in series with the switching element  22  between the input terminals  11 ,  12 , while a series circuit including the rectifier element  24  and the capacitive storage element  22  is connected in parallel with the inductive storage element  21 . In this case, the rectifier element  24  is connected in the forward direction between the capacitive storage element  26  and the inductive storage element  21 . In this buck-boost converter—in contrast to the boost converter explained above and the buck converter explained above—the output voltage Vout is of opposite polarity to the input voltage Vin. 
     The signal profiles illustrated in  FIG. 2  for the drive signal S 22  and also the first and second load path voltages V 22 , V 24  correspondingly apply to the switching converter in accordance with  FIG. 10 . 
       FIG. 11  illustrates a flyback converter as a further example of a switching converter. In the case of this flyback converter, the inductive storage element  21  is part of a transformer and forms a primary winding of the transformer, to which a secondary winding  31  is inductively coupled. The inductive storage element  21  is connected in series with the switching element  22  between the input terminals  11 ,  12 . A series circuit including the rectifier element  24  and the capacitive storage element  26  is connected in parallel with the secondary winding  31 . Optionally, a further inductive storage element  33  can be connected in the series circuit between the rectifier element  24  and the capacitive storage element  26 . In this case, a further rectifier element  32  is provided, which is then connected in parallel with a series circuit including the further inductive storage element and the capacitive storage element  26 . An output voltage Vout is present across the capacitive storage element  26 . 
     During the switched-on phase of the switching element  22 , the input voltage Vin is present across the inductive storage element  21  and the storage element  21  is magnetized. During this switched-on phase, a voltage related to the input voltage Vin by way of a turns ratio between primary winding  21  and secondary winding  31  is present across the secondary winding  31 . However, the primary winding  21  and the secondary winding of the transformer have an opposite winding sense such that, during the switched-on phase, the voltage across the secondary winding has a polarity such that the rectifier element  24  is turned off. 
     During the switched-off phase, the polarity of a voltage across the inductive storage element is reversed relative to the switched-on phase. In this case, the magnitude of the voltage is dependent on the output voltage and the turns ratio between primary and secondary windings  21 ,  31 . In this case, the maximum first load path voltage V 22   max  present across the switching element  22  during the switched-off phase corresponds to the sum of the input voltage Vin and the voltage present across the primary winding. The maximum second load path voltage V 24   max  present across the rectifier element during the switched-on phase corresponds to the voltage across the secondary winding  31  if no further inductive storage element  33  is present, and to the sum of the voltage across the secondary winding  31  and the output voltage Vout if a further inductive storage element  33  is present. 
     In the transition phase from the switched-on phase to the switched-off phase, the first load path voltage V 22  rises to the abovementioned maximum value V 22   max  while the second load path voltage V 24  decreases proceeding from the abovementioned maximum value V 24   max . During this transition phase, the first and second capacitances  23 ,  25  influence the voltage rise in the first load path voltage V 22  in the manner explained, thereby reducing for example voltage spikes which can occur at parasitic inductances in the current path with the rectifier element  24  when the rectifier element  24  accepts the inductance current I 21 . In this case, the temporal profile of the second load path voltage V 24 , which profile is influenced by the second capacitive element, influences the voltage across the secondary winding  31 . In this case, this voltage across the secondary winding  31  in turn influences the temporal profile of the voltage across the primary winding  21 , which profile in turn influences the temporal profile of the first load path voltage V 22 . In this way, the second capacitive element can influence the desired slowing down of the rate of rise of the first load path voltage V 22  toward the end of the transition phase. 
     Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.