Patent Publication Number: US-2022231610-A1

Title: Merged voltage-divider forward converter

Description:
RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 16/849,153, filed Apr. 15, 2020, which is a continuation of PCT International Application No. PCT/IB2018/057988 filed on Oct. 15, 2018, and entitled “Merged Voltage-Divider Forward Converter”, which claims priority to U.S. Provisional Patent Application No. 62/578,192 filed on Oct. 27, 2017, and entitled “Merged Capacitive-Divider Forward Converter,” all of which are hereby incorporated by reference for all purposes. 
    
    
     BACKGROUND 
     Switched mode power supplies (SMPSs) are widely used in various industrial and consumer electronic devices for the purpose of regulating an input voltage from an alternating current (AC) power source or a direct current (DC) power source such that a regulated output voltage/current may be delivered to an electronic load. 
     The circuits that make up an SMPS typically include a single-switch forward converter. Single-switch forward converter circuits are typically used when supplying loads that require tens to hundreds of watts of power with relatively high current. A single-switch forward converter circuit conveniently provides continuous output current and galvanic isolation between an input of the single-switch forward converter and an output of the single-switch forward converter. 
     In the application of the single-switch forward converter to a situation wherein a large input-to-output voltage difference is to be present, the conventional single-switch forward converter has certain limitations. These limitations may be seen to include a requirement for high voltage rated switches and a requirement for challenging transformer design. These limitations may be seen to result in a reduction of efficiency and an increase in size of the single-switch forward converter. 
     SUMMARY 
     In some embodiments, a forward converter includes an input voltage source. The input voltage source is divided into multiple divided input voltage sources. Each of the divided input voltage sources provides a portion of a total input voltage of the input voltage source. The forward converter includes an output circuit which includes a first output circuit switching device, a second output circuit switching device, an output inductor, and an output capacitor. The forward converter includes a transformer having multiple primary windings coupled to a magnetic core, a secondary winding inductively coupled to the primary windings by being coupled to the magnetic core, and a relaxation winding inductively coupled to the magnetic core. Each primary winding among the multiple primary windings is connected in series with a corresponding primary side switching device. A combination of the primary winding and the corresponding primary side switching device is in parallel with a corresponding divided voltage source among the multiple divided voltage sources. The secondary winding is connected to output an output voltage via the output circuit, and the relaxation winding is connected across the multiple divided input voltage sources or the output capacitor. The forward converter includes a controller circuit that is connected to each of the primary side switching devices and adapted to control each of the primary side switching devices in a manner that controls power flow from the input voltage source to the output capacitor, the control being based on receipt of an indication of a voltage across the output capacitor. 
     Other aspects and features of the present disclosure will become apparent to those of ordinary skill in the art upon review of the following description of specific implementations of the disclosure in conjunction with the accompanying figures. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  schematically illustrates a merged voltage-divider forward converter, where a relaxation winding is connected across a generic input voltage source, in accordance with some embodiments. 
         FIG. 2  schematically illustrates the merged voltage-divider forward converter of  FIG. 1 , where the relaxation winding is connected across an output load, in accordance with some embodiments. 
         FIG. 3  schematically illustrates a merged voltage-divider forward converter adapted to convert an AC input voltage to DC output voltage, where a relaxation winding is connected across an input voltage source, in accordance with some embodiments. 
         FIG. 4  schematically illustrates the merged voltage-divider forward converter of  FIG. 3 , where the relaxation winding is connected across an output load, in accordance with some embodiments. 
         FIG. 5  schematically illustrates a merged voltage-divider forward converter adapted to convert a DC input voltage to a DC output voltage, where a relaxation winding is connected across an input port, in accordance with some embodiments. 
         FIG. 6  schematically illustrates the merged voltage-divider forward converter of  FIG. 5 , where the relaxation winding is connected across an output load, in accordance with some embodiments. 
         FIG. 7  schematically illustrates the merged voltage-divider forward converter of  FIG. 1 , with an input source formed as a battery string and with a relaxation winding connected to the top of the battery string, in accordance with some embodiments. 
         FIG. 8  schematically illustrates the merged voltage-divider forward converter of  FIG. 2 , with an input source formed as a battery string and with a relaxation winding connected across the output load, in accordance with some embodiments. 
         FIG. 9  illustrates switching signals for primary side switches and corresponding inductor current waveforms for a continuous conduction mode of operation, in accordance with some embodiments. 
         FIG. 10  illustrates switching signals for primary side switches and corresponding inductor current waveforms for a discontinuous conduction mode of operation, in accordance with some embodiments. 
         FIG. 11  illustrates switching signals for primary side switches and corresponding inductor current waveforms for a further discontinuous conduction mode of operation, in accordance with some embodiments. 
         FIG. 12  illustrates switching signals for primary side switches and corresponding inductor current waveforms for a still further discontinuous conduction mode of operation, in accordance with some embodiments. 
         FIG. 13  illustrates simulation results from use of the forward converter in the circuit of  FIG. 6 , in accordance with some embodiments. 
         FIG. 14  illustrates a reduction of blocking voltage as a function of a number of primary windings of a transformer for the merged voltage-divider forward converter of  FIG. 5 , in accordance with some embodiments. 
         FIG. 15  illustrates a reduction of a blocking voltage as a function of a number of primary windings of a transformer for the merged voltage-divider forward converter of  FIG. 6 , in accordance with some embodiments. 
         FIG. 16  schematically illustrates the merged voltage-divider forward converter of  FIG. 2  with an additional implementation of a current sensing circuit that provides current measurement representative of current through all primary side switches and where a relaxation winding is connected across the output voltage, in accordance with some embodiments. 
         FIG. 17  schematically illustrates the merged voltage-divider forward converter of  FIG. 1  with an additional implementation of a current sensing circuit that provides current measurement representative of current through all primary switches and through a relaxation winding, where the relaxation winding is connected across a generic input voltage source, in accordance with some embodiments. 
         FIG. 18  schematically illustrates the merged voltage-divider forward converter of  FIG. 5  with an additional implementation of a snubber circuit including a snubber resistor, a snubber diode and a snubber capacitor, in accordance with some embodiments. 
         FIG. 19  schematically illustrates the merged voltage-divider forward converter of  FIG. 6  with an additional implementation of a snubber circuit including a snubber resistor, a snubber diode and a snubber capacitor, in accordance with some embodiments. 
         FIG. 20  schematically illustrates the merged voltage-divider forward converter of  FIG. 5  with an additional implementation of a snubber circuit including a snubber diode, a snubber Zener diode and a snubber capacitor, in accordance with some embodiments. 
         FIG. 21  schematically illustrates the merged voltage-divider forward converter of  FIG. 6  with an additional implementation of a snubber circuit including a snubber diode, a snubber Zener diode and a snubber capacitor, in accordance with some embodiments. 
         FIG. 22  schematically illustrates the merged voltage-divider forward converter of  FIG. 5  with an additional implementation of an active snubber circuit including an active snubber switch and a snubber capacitor, in accordance with some embodiments. 
         FIG. 23  schematically illustrates the merged voltage-divider forward converter of  FIG. 6  with an additional implementation of an active snubber circuit including an active snubber switch and a snubber capacitor, in accordance with some embodiments. 
     
    
    
     DETAILED DESCRIPTION 
     Some embodiments disclosed herein involve a merged voltage-divider forward converter that is coupled to a series combination of divided input voltage sources to provide output power to a load. In some embodiments, the series combination of divided input voltage sources includes a series combination of capacitors. An input voltage source is coupled across the series combination of capacitors and each capacitor acts as a divided input voltage source to provide a divided voltage level, i.e., a portion, of the total input voltage to the forward converter. In other embodiments, the series combination of energy sources includes a series combination of battery cells. Each battery cell provides a divided voltage level of the total input voltage provided to the forward converter. 
     An input stage of the forward converter is galvanically isolated from an output stage of the forward converter by a transformer. As disclosed herein, the transformer has multiple primary windings and each of the primary windings is coupled to a respective divided voltage source of the series combination of divided input voltage sources. Each of the primary windings is further coupled in a series combination with a respective primary side switching device to control a current through the respective primary winding. Each series combination is coupled in parallel across a respective divided input voltage source. Due to the smaller, divided input voltages of either of the above embodiments (i.e., a capacitive divider or a string of battery cells), both a dimensional volume of the transformer and a power efficiency of the transformer can be optimized. The reduced volume of the transformer can significantly reduce the volume of the forward converters disclosed herein. Such optimization is achieved by reducing a voltage swing across the transformer magnetizing inductance and across an output inductor of the forward converter. 
     Additionally, due to the smaller, divided input voltages of either of the above embodiments (i.e., a capacitive divider or a string of battery cells), the switching voltage of the primary side switching devices may be advantageously reduced. The reduced volt-seconds across magnetic elements and reduced switching voltages advantageously allow for use of reduced-size magnetic components relative to conventional designs. Conveniently, use of reduced-size magnetic components and lower switching voltages results in lower magnetic and switching losses, based on principles known in the field of multi-level converters. More specifically, the volume of the transformer can be minimized, thereby allowing for the use of low-voltage switching devices for the primary side switching devices. Typically, such low-voltage switching devices can be MOSFETs, bipolar junction transistors (BJTs) or insulated-gate bipolar transistors (IGBTs). Moreover, low-voltage MOSFETs are desirable for use as the primary side switching devices due to lower switching losses, lower conduction losses and lower cost associated with MOSFETs relative to BJTs and IGBTs. 
     Additionally, the forward converters disclosed herein advantageously reduce a peak magnitude of the input current while also lowering frequency harmonics of the input current. Such forward converters distribute input power losses over the multiple primary windings and primary side switching devices. Because of the reduced magnitude of the peak input current and because of the elimination of lower frequency harmonics of the input current, a corner frequency of an (EMI) input filter described herein (with reference to  FIG. 3  and  FIG. 4 ) can be increased, potentially reducing the volume of the input filter. 
     Further, a passive balancing of the input voltage across the divided input voltage sources can be obtained. For example, when a relatively small voltage misbalance condition occurs amongst the divided input voltage sources, one or more body diodes of the primary side switching devices can become forward biased, thus enabling electric charge to be redistributed from the capacitive device/battery cell with the largest voltage to other capacitive devices/battery cells with lower voltages. An equivalent circuit may be considered to have been formed during the charge redistribution. The equivalent circuit is an LC type, consisting of the input capacitances and leakage inductances. Such an LC type equivalent circuit may be seen to limit peak current magnitude. Thus, a maximum passive operation input voltage balancing can be obtained. Notably, the maximum passive operation input voltage misbalance may be shown to be less than the forward voltage of the body diode of the primary side switching device  160  (&lt;1.2 volts usually). 
       FIG. 1  illustrates a forward converter  100  connected to receive an input voltage, v in (t), from a generic input voltage source  110 , and output an output voltage, v out , to an output load  150 , in accordance with some embodiments. The generic input voltage source  110  is representative of either an AC voltage source or a DC voltage source. 
     The forward converter  100  includes a transformer  130  with a number, N, of primary windings including, for example, a first primary winding  132 - 1 , a second primary winding  132 - 2 , and an N th  primary winding  132 -N. The primary windings may be, individually or collectively, associated with reference numeral  132 . A particular one of the primary windings  132  may be associated with reference numeral  132 - k , where k is an integer selected from an array including integers from 1 to N, where N is the number of primary windings  132 . The first primary winding  132 - 1  has a number of turns represented by n p,1 ; the second primary winding  132 - 2  has a number of turns represented by n p,2 , and the N th  primary winding  132 -N has a number of turns represented by n p,N . 
     The first primary winding  132 - 1  is inductively coupled to a secondary winding  134  of the transformer  130 . The secondary winding  134  has a number of turns represented by n s . A first node of the secondary winding  134  is connected to an anode of a first diode  142  (D 1 ) (i.e., a first output switching device), and a second node of the secondary winding  134  is connected to an anode of a second diode  144  (D 2 ) (i.e., a second output switching device). The cathodes of the diodes  142 ,  144  are connected to each other and to one terminal of an output inductor  146  (with a value represented by L d ). Though diodes  142 ,  144  are used to implement the particular embodiment of  FIG. 1 , each of the diodes  142 ,  144  may be understood to represent a device that is more generically a switch that operates as voltage-unidirectional and current-unidirectional. For example, in some embodiments, one or both of the diodes  142 ,  144  may instead be implemented as a MOSFET under active control. 
     The other terminal of the output inductor  146  is connected to a terminal of an output capacitor  148  (with a value represented by C out ) and to a terminal of the output load  150  to provide an output voltage v out  to the output load  150 . The other terminal of the output capacitor  148  is connected to the other terminal of the output load  150  and to the anode of the second diode  144 . 
     The generic input voltage source  110  is connected to a capacitive divider  120  which divides the generic input voltage source  110  into a series of divided input voltage sources, each capacitive device of the capacitive divider  120  being one of the divided input voltage sources. The capacitive divider  120  has a first capacitive device  122 - 1  (with value C in;1 ), a second capacitive device  122 - 2  (with value C in;2 ), and an N th  capacitive device  122 -N (with value C in;N ) connected in series. The capacitive devices may be, individually or collectively, associated with reference numeral  122 . A particular one of the capacitive devices may be associated with reference numeral  122 - k , where k is an integer selected from an array including integers from 1 to N, where N is the number of primary windings  132  and the number of capacitive devices  122 . The capacitive device  122 - k  has a respective capacitance value C in;k  associated therewith. In operation, the generic input voltage source  110  powers the forward converter  100  with an input voltage, v in (t). The capacitive divider  120  receives the input voltage v in (t). Along the capacitive divider  120 , each one of the capacitive devices  122  provides a smaller, divided voltage V in;k . 
     The forward converter  100  includes N primary side switching devices SW 1    160 - 1  through SW N    160 -N.  FIG. 1  shows a first primary side switching device SW 1    160 - 1 , a second primary side switching device SW 2    160 - 2 , and an N th  primary side switching device SW N    160 -N. The primary side switching devices  160 - 1  through  160 -N may be, individually or collectively, associated with reference numeral  160 . A particular one of the primary side switching devices  160  may be associated with reference numeral  160 - k , where k is an integer selected from an array including integers from 1 to N, where N is the number of primary windings  132 , the number of capacitive devices  122 , and the number of primary side switching devices  160 . The primary side switching devices  160  can be advantageously provided in the form of metal-oxide-semiconductor field-effect transistors (MOSFETs), which can manage the relatively small divider voltages V in;k . 
     Each primary winding  132 - k  is connected to a corresponding primary side switching device  160 - k  in a series combination. Each of these series combinations of a primary winding  132 - k  and corresponding primary side switching device  160 - k  is connected in parallel across a corresponding capacitive device  122 - k  of the capacitive divider  120 . For example, the first primary winding  132 - 1  is connected in a first series combination with the first primary side switching device  160 - 1 , the first series combination being connected in parallel across the first capacitive device  122 - 1 ; the second primary winding  132 - 2  is connected in a second series combination with the second primary side switching device  160 - 2 , the second series combination being connected in parallel across the second capacitive device  122 - 1 , and the N th  primary winding  132 -N is connected in an N th  series combination with the N th  primary side switching device  160 -N, the N th  series combination being connected in parallel across the N th  capacitive device  122 -N. 
     The primary side switching devices  160  are operatively controlled by a controller circuit  170 . In particular, the controller circuit  170  provides a first control signal c 1  to the first primary side switching device  160 - 1 , provides a second control signal c 2  to the primary side second switching device  160 - 2 , and provides an N th  control signal c N  to the N th  primary side switching device  160 -N. 
     When one of the primary side switching devices  160 - k  is enabled (e.g., conducting current), the corresponding primary winding  132 - k  may be characterized by a magnetizing inductance value L m  illustrated as a representative inductor  138 , with value L m  that is connected in parallel with the N th  primary winding  132 -N. The magnetizing inductance value L m  may vary with a dependence on which of the primary side switching devices  160  is enabled. 
     An output voltage conditioner  152  receives the output voltage, v out (t), and conditions the output voltage for use by the controller circuit  170 . The output voltage conditioner  152  may, for example, condition the output voltage through the use of a combination of a resistive voltage divider (e.g., implementing an 
     
       
         
           
             
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     transfer function, not shown) and a low-pass, 1 st  order, RC filter (not shown). Thus, the output voltage conditioner  152  conditioning may act to step-down the output voltage and filter the output voltage before the conditioned output voltage is received by the controller circuit  170 . 
     The forward converter  100  of  FIG. 1  is configured to receive the input voltage v in (t) at the capacitive divider  120  and to successively and repeatedly operate, using the primary side switching devices  160 , each one of the primary windings  132  using a portion, V in;k , of the input voltage via the corresponding capacitive device  122  to generate the output voltage v out  to the output load  150  of the forward converter  100 . It is noted that the operation of the primary side switching devices  160  during use of the forward converter  100  can be performed in any sequential or logical order. 
     In operation, the controller circuit  170  may base control (through control signals c 1 , c 2 , . . . , C N ) of the primary side switching devices  160  upon receiving, from the output voltage conditioner  152 , a conditioned indication of a voltage across the output capacitor  148 . Each one of the primary side switching devices  160  is operatively controlled by the controller circuit  170  for operating the forward converter  100  of  FIG. 1  successively and repeatedly between an on-state (i.e., enabled) and an off-state (i.e., disabled). Advantageously, during normal use of the forward converter  100 , only one primary side switching device  160  is operated at a time, thereby enabling distribution of power losses over all input stage components. Such distribution of power losses may be seen to act to minimize hot spots, thus reducing cooling requirements. 
     The forward converter  100  includes a relaxation winding  136  connected in series with a relaxation diode  162  (D r ). The relaxation winding  136  has a number of turns represented by n r . Though the relaxation diode  162  is implemented as a diode in the embodiment shown, the relaxation diode  162  may be understood to represent a device that is more generically a switch that operates as voltage-unidirectional and current-unidirectional. For example, in some embodiments, the relaxation diode  162  may be implemented as a MOSFET under active control. In the embodiment shown in  FIG. 1 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected across and in parallel to the capacitive divider  120 , and thus also across terminals of the generic input voltage source  110 . Thus, each serial combination of a primary winding  132  and corresponding primary side switching device  160  is not coupled to a unique relaxation winding. In particular, one or more of the primary windings  132  (e.g.,  132 - 2 ) are not directly connected to the relaxation winding  136 . Further, one or more of the primary side switching devices  160  are not directly connected to the relaxation diode  162  and therefore are not directly coupled to the relaxation winding  136  through the relaxation diode  162 . Further, though the first primary winding  132 - 1  is directly connected to the relaxation winding  136 , the first primary side switching device  162 - 1  is not directly connected to the relaxation diode  162 . Rather, the N th  primary side switching device  160 -N is directly connected to the relaxation diode  162 . 
     Connecting a single relaxation winding (the relaxation winding  136 ) across the full input voltage source (the generic voltage input source  110 ), as opposed to a stacking of the forward converter primary structure which requires having individual relaxation windings and diodes for each primary structure of the stack, or having a relaxation winding across only one of the N primary windings, has multiple benefits. Compared to a multi-relaxation winding solution, the embodiment shown in  FIG. 1  reduces the number of components required (e.g., diodes and windings) for relaxation by N−1 as compared to having N relaxation windings and diodes. Secondly, the embodiment shown in  FIG. 1  helps achieve charge balance across the capacitive divider devices  122  and therefore achieves higher efficiency as compared to the aforementioned other two methods. This is because the magnetizing inductance energy of the forward converter  100  will charge the series combination of those capacitive devices  122 . Thirdly, the embodiment shown in  FIG. 1  reduces the number of winding turns required for relaxation winding by a factor of N as compared to a conventional forward converter to achieve similar performance and maximum allowable duty cycle (where the max duty cycle is 
     
       
         
           
             
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     for a forward converter or the other two methods mentioned that can be used to connect relaxation winding/windings on the primary side). 
     In other embodiments, the series combination of the relaxation winding  136  and the relaxation diode  162  is connected to a terminal of an output capacitor  148 . For example,  FIG. 2 , illustrates a forward converter  200  which is similar to the forward converter  100  of  FIG. 1 , but the forward converter  200  is configured such that the series combination of the relaxation winding  136  and the relaxation diode  162  are connected across the terminals of the output capacitor  148 . Otherwise, aspects of the forward converter  200  are similar to the forward converter  100 . That is, the forward converter  200  includes the generic input voltage source  110 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . The embodiment shown in  FIG. 2  operates based on a different principle as compared to a conventional forward converter. That is, the magnetizing inductance  138  of the transformer  130  is advantageously used as a means of energy transfer to the output through the relaxation winding  136  as opposed to being a parasitic component generating extra loss. Therefore, opposite to a conventional forward converter where an ideal transformer with very high magnetizing inductance is needed to minimize magnetizing current of its magnetizing inductance  138  and therefore core losses, as well as resistive losses through components that carry the magnetizing inductance  138  current (namely, primary side semiconductor devices and primary side transformer windings), the transformer  130  can advantageously be designed to have much smaller magnetizing inductance  138  by up to three orders of magnitude therefore requiring significantly smaller number of winding turns, and thereby reducing transformer volume. This is enabled by the fact that the energy stored in the magnetizing inductance  138  and current going through it is used to transfer power to the output rather than being sent back and forth between V in  and the magnetizing inductance  138  and eventually being dissipated as heat in methods where the relaxation winding  136  connects to the input. This transformer size reduction due to different principle of operation comes in addition to the transformer size reduction due to using multiple primary windings that reduces input voltage to the primary windings  132  by a factor of N and thereby reducing volt-seconds across the magnetic core, and also the reduced transformer division ratio for step-down applications. In addition, the transformer  130  can be advantageously designed with an air gap similar to that of a flyback transformer (which in principle is a coupled inductor where there magnetizing inductance  138  is used for power transfer rather than a true transformer that is needed by a conventional forward converter) to allow for higher saturation current. 
       FIG. 3  and  FIG. 4  illustrate example embodiments of forward converters  300 ,  400  configured to receive an AC voltage, in accordance with some embodiments. The forward converter  300  is similar to the forward converter  100 , and the forward converter  400  is similar to the forward converter  200 . That is, each of the forward converters  300 ,  400  includes the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . As illustrated in  FIG. 3  and  FIG. 4 , when configured to receive an AC voltage, the generic input voltage source  110  may include an AC input voltage source  190 , an electromagnetic interference (EMI) input filter  192 , and a full-wave diode rectifier  194 . The forward converter  300  of  FIG. 3  and  FIG. 4  has the EMI input filter  192  operatively connected to the AC input voltage source  190  for providing a filtered input voltage either directly or indirectly to the capacitive divider  120 . In the specific embodiment illustrated in  FIGS. 3 and 4 , the EMI input filter  192  is indirectly connected to the capacitive divider  120  via the full-wave diode rectifier  194 . The full-wave diode rectifier  194  is generally provided to rectify the signal provided by the AC input voltage source  190 . 
     In  FIG. 3 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIG. 1 , across input terminals of the forward converter  300  (across terminals of the full-wave diode rectifier  194 ). In  FIG. 4 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIG. 2 , across terminals of the output capacitor  148 . 
     In both  FIG. 3  and  FIG. 4 , the primary side MOSFET switching devices  160  (of  FIGS. 1 and 2 ) are replaced with more general primary side single-pole, single throw switching devices. That is, the first MOSFET primary side switching device  160 - 1  (in  FIGS. 1 and 2 ) is replaced (in  FIGS. 3 and 4 ) with a first primary side single-pole, single throw switch  161 - 1 . Further, the second primary side MOSFET switching device  160 - 2  (in  FIGS. 1 and 2 ) is replaced (in  FIGS. 3 and 4 ) with a second primary side single-pole, single throw switch  161 - 2 . Still further, the N th  primary side MOSFET switching device  160 -N (in  FIGS. 1 and 2 ) is replaced (in  FIGS. 3 and 4 ) with an N th  primary side single-pole, single throw switch  161 -N. This further illustrates that for each of the embodiments disclosed herein, the primary side switching devices  160  can be more generally described as a switch which can include a FET, a MOSFET, a BJT, an IGBT, and so on. 
       FIG. 5  and  FIG. 6  illustrate example embodiments of forward converters  500 ,  600  configured to receive a DC voltage input, in accordance with some embodiments. The forward converter  500  is similar to the forward converter  100 , and the forward converter  600  is similar to the forward converter  200 . That is, each of the forward converters  500 ,  600  includes the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . As illustrated in  FIG. 5  and  FIG. 6 , when configured for DC voltage input, the generic input voltage source  110  may be realized as a DC input voltage source  180 . The forward converter  500  and the forward converter  600  are each connected between the DC input voltage source  180  and the output load  150  in accordance with aspects of the present application. In  FIG. 5 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIG. 1  and  FIG. 3 , across input terminals of the forward converter  500 , that is, across terminals of the DC input voltage source  180 . In  FIG. 6 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIG. 2  and  FIG. 4 , across terminals of the output capacitor  148 . 
       FIG. 7  and  FIG. 8  illustrate additional example embodiments of forward converters  700 ,  800  configured to receive a DC voltage input, in accordance with some embodiments. The forward converter  700  is similar to the forward converter  100 , and the forward converter  800  is similar to the forward converter  200 . That is, each of the forward converters  700 ,  800  includes the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . As illustrated in  FIG. 7  and  FIG. 8 , when configured for DC voltage input, the generic input voltage source  110  can be realized as an input voltage source string. In the embodiment shown, the input voltage source string is realized as a battery string  410 , each battery of the battery string  410  providing a divided input voltage. The battery string  410  may be implemented by N batteries in series combination, the series combination including: a first battery  410 - 1 ; a second battery  410 - 2 ; and an N th  battery  410 -N. A particular one of the batteries of the battery string  410  may be associated with reference numeral  410 - k , where k is an integer selected from an array including integers from 1 to N, where N is the number of primary windings  132 . Each battery  410 - k  provides a divided input voltage V in;k . For example, the battery  410 - 1  provides a voltage V in;1 , the battery  410 - 2  provides a voltage V in;2 , and the battery  410 -N provides a voltage V in;N . As shown, each series combination of a primary winding  132 - k  and corresponding primary side switching device  160 - k  is connected in parallel across a corresponding battery  410 - k . In  FIG. 7 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIGS. 1, 3 and 5 , across the input terminals of the forward converter  700 , now the top of the battery string  410 . In  FIG. 8 , the series combination of the relaxation winding  136  and the relaxation diode  162  is connected, as in  FIGS. 2, 4 and 6 , across the terminals of the output capacitor  148 . 
       FIG. 9  illustrates simplified example switching signals for primary side switching devices  160  and corresponding inductor current waveforms in response to control signals c 1  through c N  generated by the controller circuit  170 , in accordance with some embodiments where the series combination of the relaxation winding  136  and the relaxation diode  162  is connected across the terminals of the output capacitor  148 . A waveform c includes a series of pulses corresponding to an aggregate of control signals c 1  through c N . That is, each pulse of waveform c corresponds to a pulse of one or more control signal c 1  through c N . The switching waveforms of  FIG. 9  depict a Continuous Conduction Mode (CCM) of operation for the case when output inductor  146  current i L  is continuous and magnetizing inductance  138  current i LM  is continuous. That is, each of the currents i L  and i LM  maintain a value greater than 0. 
     When the first primary side switching device  160 - 1  is enabled responsive to a high value of control signal c 1 , the first diode  142  is positively biased and conducts the current, while the second diode  144  and the relaxation diode  162  are reverse-biased. During this time, a current, i Lm , through the magnetizing inductance  138 , ramps up to an i Lm  peak  904 . The i Lm  peak  904  is referred to herein as I Lmp . Furthermore, the current, i L , through the output inductor  146  ramps up, as well, to an i L  peak  902 . The i L  peak  902  is referred to herein as I p . The i L  peak  902  illustrates the mode of operation where the output inductor  146  is operated in CCM mode. Furthermore, an i 1  peak  906  for the current, i 1 , through the first primary winding  132 - 1  may be determined from 
     
       
         
           
             
               
                 
                   n 
                   
                     p 
                     ; 
                     N 
                   
                 
                 
                   n 
                   
                     p 
                     ; 
                     1 
                   
                 
               
               ⁢ 
               
                 I 
                 Lmp 
               
             
             + 
             
               
                 
                   n 
                   s 
                 
                 
                   n 
                   
                     p 
                     ; 
                     1 
                   
                 
               
               ⁢ 
               
                 
                   I 
                   p 
                 
                 . 
               
             
           
         
       
     
     Additionally, an i 2  peak  908  for the current, i 2 , through the second primary winding  132 - 2  may be determined from 
     
       
         
           
             
               
                 
                   
                     n 
                     
                       p 
                       ; 
                       N 
                     
                   
                   
                     n 
                     
                       p 
                       ; 
                       2 
                     
                   
                 
                 ⁢ 
                 
                   I 
                   Lmp 
                 
               
               + 
               
                 
                   
                     n 
                     s 
                   
                   
                     n 
                     
                       p 
                       ; 
                       2 
                     
                   
                 
                 ⁢ 
                 
                   I 
                   p 
                 
               
             
             , 
           
         
       
     
     and an i N  peak  910  for the current, i N , through the N th  primary winding  132 -N may be determined from 
     
       
         
           
             
               I 
               Lmp 
             
             + 
             
               
                 
                   n 
                   s 
                 
                 
                   n 
                   
                     p 
                     ; 
                     N 
                   
                 
               
               ⁢ 
               
                 
                   I 
                   p 
                 
                 . 
               
             
           
         
       
     
     When none of the primary side switching devices  160  are enabled, the first diode  142  is reverse-biased, i.e., the first diode  142  is in the off-state. This event corresponds to the low value of the control signal c 1 , c 2 , . . . , c N  of  FIG. 9 . In this case, as long as the inductor current, i L , is still positive, the second diode  144  is positively biased, i.e., the second diode  144  is in an on-state. Additionally, as long as the magnetizing inductor current, i Lm , is still positive, the relaxation diode  162  is positively biased, i.e., the relaxation diode  162  is also in the on-state. The second diode  144  conducts the current, i L , that flows through the output inductor  146 , while the relaxation diode  162  conducts the current, i Lm , through the magnetizing inductance  138 . During this time interval, as shown in  FIG. 9 , the current, i Lm , through the magnetizing inductance  138  ramps down, and the current, i L , of the output inductor  146  ramps down as well. 
     In accordance with the embodiment illustrated in  FIG. 1 , the magnetizing current, i Lm , flows back to the generic input voltage source  110  when the relaxation diode  162  is in the on-state. In the embodiment illustrated in  FIG. 2 , the magnetizing current, i Lm , flows to the output load  150  when the relaxation diode  162  is in the on-state. 
     Switching waveforms represented in  FIGS. 10, 11 and 12  depict various Discontinuous Conduction Modes (DCMs) of operation. The DCM of operation is a name for a mode wherein either one of the magnetizing current, i Lm , or the output inductor current, i L , is discontinuous. Similarly, a Boundary Conduction Mode (BCM) of operation may be achieved by operating either of the two magnetizing or output inductor currents at the boundary between the CCM of operation and the DCM of operation. A Quasi Resonant (QR) mode of operation may be achieved if the turn-on time of the primary side switching devices  160 ,  161  is delayed in the BCM of operation such that the switch is turned on at the instant where either the voltage across the magnetizing inductance  138  or the voltage across the output inductor  146  reaches a minimum after the respective current goes to zero. 
       FIG. 10  illustrates simplified example switching signals for primary side switching devices  160 / 161  and corresponding inductor current waveforms in response to control signals c 1  through c N  generated by the controller circuit  170 , in accordance with some embodiments where the series combination of the relaxation winding  136  and the relaxation diode  162  is connected across the terminals of the output capacitor  148 . In  FIG. 10 , the current, i Lm , through the magnetizing inductance  138  ramps up to an i Lm  peak, I Lmp , represented by reference numeral  1004 . Furthermore, the current, i L , through the output inductor  146  ramps up, as well, to an i L  peak, I p , represented by reference numeral  1002 . I p    1002  represents a DCM operating mode for the output inductor  146  where the current i L  must reach zero at every switching cycle during steady state operation. Furthermore, an i 1  peak  1006  for the current, i 1 , through the first primary winding  132 - 1 , an i 2  peak  1008  for the current, i 2 , through the second primary winding  132 - 2  and an i N  peak  1010  for the current, i N , through the N th  primary winding  132 -N may be determined in the same manner as described hereinbefore with reference to the peaks in  FIG. 9 . 
       FIG. 11  illustrates simplified example switching signals for primary side switching devices  160  and corresponding inductor current waveforms in response to control signals c 1  through c N  generated by the controller circuit  170 , in accordance with some embodiments where the series combination of the relaxation winding  136  and the relaxation diode  162  is either connected across embodiments of the generic input voltage source  110  or connected across the terminals of the output capacitor  148 . In  FIG. 11 , the current, i Lm , through the magnetizing inductance  138  ramps up to an i Lm  peak, I Lmp , represented by reference numeral  1104 . I Lmp    1104  represents a DCM operating mode for the magnetizing inductance L m    138  where current must reach zero at every switching cycle during steady state operation. Furthermore, the current, i L , through the output inductor  146  ramps up, as well, to an i L , peak, I p , represented by reference numeral  1102 . Furthermore, an i 1  peak  1106  for the current, i 1 , through the first primary winding  132 - 1 , an i 2  peak  1108  for the current, i 2 , through the second primary winding  132 - 2  and an i N  peak  1110  for the current, i N , through the N th  primary winding  132 -N may be determined in the same manner as described hereinbefore with reference to the peaks in  FIG. 9 . 
     As shown in  FIGS. 9-11 , the magnetizing inductance  138  can be operated in DCM (discontinuous conduction mode), QR (quasi-resonant mode), BCM (boundary conduction mode), or CCM (continuous conduction mode). The choice of mode of operation depends on the application, input-to-output voltage ratio, load current and overall design optimization for the output power delivery division and sizing of i) a buck-based circuit formed by the secondary winding  134 , the diodes  142 ,  144 , and the output inductor  146 , and ii) a buck-boost-based circuit formed by the relaxation winding  136  and the relaxation diode  162 . This advantageous flexibility is contrary to how a conventional forward converter operates, whereby the magnetizing inductance of a transformer of the conventional forward converter has to be operated only in DCM mode to avoid core saturation of the transformer and to minimize conduction losses. 
     In a conventional forward converter, a converter primary side switching device is controlled similar to a buck converter with respect to a secondary side rectifier device, and an output voltage DC level of the conventional forward converter is controlled by duty cycle similar to a buck converter with additional multiplication factor of 
     
       
         
           
             
               n 
               s 
             
             
               n 
               p 
             
           
         
       
     
     due to transformer turns ratio. Thus, the output DC voltage level of the conventional forward converter is governed by the equation: 
     
       
         
           
             
               
                 v 
                 
                   o 
                   ⁢ 
                   u 
                   ⁢ 
                   t 
                 
               
               = 
               
                 
                   D 
                   ⁢ 
                   
                     n 
                     s 
                   
                 
                 
                   n 
                   p 
                 
               
             
             , 
           
         
       
     
     where D is the duty cycle when operated in BCM or CCM mode. The buck-based power stage can be operated in DCM, BCM or CCM mode, but with a limitation on the duty cycle. However, in order to avoid core saturation of the transformer, a magnetizing inductance of a transformer of the conventional forward converter must be guaranteed by design to operate only in DCM mode, resulting in a limitation on duty cycle of the conventional forward converter (as seen by the output inductor of the forward converter) to be smaller than 
     
       
         
           
             
               
                 n 
                 
                   p 
                   , 
                   i 
                 
               
               
                 
                   n 
                   
                     p 
                     , 
                     i 
                   
                 
                 + 
                 
                   n 
                   r 
                 
               
             
             . 
           
         
       
     
     That is, the buck-based circuit refers to the circuit components that perform power transfer through the output inductor  146 . The output inductor  146  can be operated in DCM, BCM, or CCM modes. The buck-boost-based circuit refers to the circuit that performs power transfer through the magnetizing inductance L m    138 . In the case of conventional forward converter, the magnetizing inductance L m  has to be operated in DCM mode (i.e., the current through the magnetizing inductance L m  must go to zero at every cycle). 
     In some embodiments of forward converters disclosed herein, the principle of operation and therefore control mechanism is different from that of a conventional forward converter as described below. That is, forward converters disclosed herein which have a relaxation winding on the secondary side (e.g., the forward converters  200 ,  400 ,  600 ,  800 ,  1600 ,  1900 ,  2100 ,  2300 ) have a multi-level, series connection, primary-side stage which is formed by the primary side switching devices  160 / 161  and primary side windings  132 . This primary-side stage is shared and simultaneously used by two parallel circuits that deliver power to the load, the parallel circuits being i) the buck-based circuit formed by the secondary winding  134 , the diodes  142 ,  144 , and the output inductor  146 , and ii) the buck-boost-based circuit formed by the relaxation winding  136  and the relaxation diode  162 . Thus, such forward converters can be said to have a serial input, parallel output topology. As previously explained, the controller circuit  170  can be configured to operate each of the parallel outputs in CCM, BCM, or DCM. The buck-boost based circuit formed by the relaxation winding  136  and the relaxation diode  162  can operate in QR mode as well to reduce switching losses for the primary side switching devices  160 / 161  at its turn-on transition and therefore minimize switching losses. The series connected input allows for reduced semiconductor voltage ratings, reduced switching frequency for each primary side device, distribution of heat across multiple devices and transformer size reduction as described in this disclosure. 
     When the buck-based output (output inductor  146  current i L ) is operated in BCM or DCM mode, the output voltage DC is controlled and governed by the equation: 
     
       
         
           
             
               
                 v 
                 
                   o 
                   ⁢ 
                   u 
                   ⁢ 
                   t 
                 
               
               = 
               
                 
                   v 
                   
                     i 
                     ⁢ 
                     n 
                   
                 
                 ⁢ 
                 
                   
                     D 
                     ⁢ 
                     
                       n 
                       s 
                     
                   
                   
                     N 
                     ⁢ 
                     
                       n 
                       
                         p 
                         , 
                         i 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
     where D is the duty cycle seen by the output inductor  146 . When the magnetizing inductance  138  is operated in BCM or CCM, the output voltage v out  DC level is controlled and governed by the equation: 
     
       
         
           
             
               
                 v 
                 
                   o 
                   ⁢ 
                   u 
                   ⁢ 
                   t 
                 
               
               = 
               
                 
                   v 
                   
                     i 
                     ⁢ 
                     n 
                   
                 
                 ⁢ 
                 
                   
                     D 
                     ⁢ 
                     
                       n 
                       r 
                     
                   
                   
                     
                       ND 
                       ′ 
                     
                     ⁢ 
                     
                       n 
                       
                         p 
                         , 
                         i 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
     where D is the duty cycle seen by the magnetizing inductance  138  and D′ is a primary side switching device off-time duty cycle, equal to 1-D. Therefore, by designing the transformer winding ratios for a given V in  and V out , each or both of the two inductors  138 ,  146  can be pushed into CCM to deliver higher load current as shown in  FIGS. 9-11 . This choice depends on the design of the transformer  130 , the output inductor  146  and the selection of the diodes  142 ,  144 ,  162  and primary side switching devices  160 / 161  to achieve maximum efficiency and minimized volume for a specific application with given load current and input and output voltages. The determining factor for mode of operation is 
     
       
         
           
             
               
                 n 
                 r 
               
               
                 
                   n 
                   s 
                 
                 ⁢ 
                 
                   D 
                   ′ 
                 
               
             
             , 
             
               
                 where 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 
                   
                     n 
                     r 
                   
                   
                     
                       n 
                       s 
                     
                     ⁢ 
                     
                       D 
                       ′ 
                     
                   
                 
               
               &lt; 
               1 
             
           
         
       
     
     results in the output inductor  146  (buck-based output) being pushed to CCM before the magnetizing inductance  138  (buck-boost-based output), where the magnetizing inductance  138  is operated in DCM during steady-state operation and vice versa for the case 
     
       
         
           
             
               
                 n 
                 r 
               
               
                 
                   n 
                   s 
                 
                 ⁢ 
                 
                   D 
                   ′ 
                 
               
             
             &gt; 
             1. 
           
         
       
     
     Therefore, having the known n r , n s  and duty cycle, the controller circuit  170  selects the mode of operation to maximize efficiency, and in order to optimize the dynamic performance, the controller circuit  170  can switch between different dynamic models at the time of mode switching. 
     For operating the magnetizing inductance  138  in DCM mode of operation, the criteria for the described embodiment herein is 
     
       
         
           
             D 
             &lt; 
             
               
                 N 
                 ⁢ 
                 
                   n 
                   
                     p 
                     , 
                     i 
                   
                 
                 ⁢ 
                 
                   v 
                   
                     o 
                     ⁢ 
                     u 
                     ⁢ 
                     t 
                   
                 
               
               
                 
                   
                     n 
                     r 
                   
                   ⁢ 
                   
                     v 
                     
                       i 
                       ⁢ 
                       n 
                     
                   
                 
                 + 
                 
                   N 
                   ⁢ 
                   
                     n 
                     
                       p 
                       , 
                       i 
                     
                   
                   ⁢ 
                   
                     v 
                     
                       o 
                       ⁢ 
                       u 
                       ⁢ 
                       t 
                     
                   
                 
               
             
           
         
       
     
     whereas for a conventional forward converter, 
     
       
         
           
             
               D 
               ⁢ 
               
                   
               
               ⁢ 
               is 
             
             &lt; 
             
               
                 
                   n 
                   
                     p 
                     , 
                     i 
                   
                 
                 
                   
                     n 
                     
                       p 
                       , 
                       i 
                     
                   
                   + 
                   
                     n 
                     r 
                   
                 
               
               . 
             
           
         
       
     
     However, as explained, DCM operation for the magnetizing inductance  138  is not required for the embodiments of forward converter disclosed herein as opposed to conventional forward converters. 
       FIG. 12  illustrates simplified example switching signals for primary side switching devices  160 / 161  and corresponding inductor current waveforms in response to control signals ci through c N  generated by the controller circuit  170 , in accordance with some embodiments where the series combination of the relaxation winding  136  and the relaxation diode  162  is either connected across embodiments of the generic input voltage source  110  or connected across the terminals of the output capacitor  148 . In  FIG. 12 , the current, i Lm , through the magnetizing inductance  138  ramps up to an i Lm  peak, I Lmp , represented by reference numeral  1204 . I Lmp    1204  corresponds to a DCM operating mode for the magnetizing inductance L m    138  where current must reach zero at every switching cycle during steady state operation. Furthermore, the current, i L , through the output inductor  146  ramps up, as well, to an i L , peak, I p , represented by reference numeral  1202 . Furthermore, an i 1  peak  1206  for the current, i 1 , through the first primary winding  132 - 1 , an i 2  peak  1208  for the current, i 2 , through the second primary winding  132 - 2  and an i N  peak  1210  for the current, i N , through the N th  primary winding  132 -N may be determined in the same manner as described hereinbefore with reference to the peaks in  FIG. 9 . 
       FIG. 13  illustrates plots representative of example simulation results of use of the forward converter  600  of  FIG. 6 , in accordance with some embodiments, where two primary windings are used (N=2) with the following parameters: V ds =400 V; v out =20 V; I load =8 A; L m =60 μH; L d =33 μH; n p;1 =6; n p;2 =6; n s =5; n r =3. The first two waveforms of  FIG. 13  show the control (gate driving) signals c 1  and c 2  of the primary side switching devices  160 - 1  and  160 - 2  when the forward converter  600  operates in accordance with the waveforms shown in  FIG. 11  (i.e., the output inductor  146  (of the buck-based power stage) is operated in CCM mode and the magnetizing inductance L m    138  (of the buck-boost based power stage) is operated in DCM mode). The first diode  142  conducts current i L  of the output inductor  146  when one of the primary side switching devices  160  is enabled. When all of the primary side switching devices  160  are disabled, the current i L  goes through the second diode  144 . In  FIG. 13 , the magnetizing inductance L m    138  is operated in DCM mode. When one of the primary side switching devices  160  is enabled, the magnetizing inductance L m    138  is charged. When all of the primary side switching devices  160  are disabled, the current i Lm  conducts through the relaxation winding  136  and the relaxation diode D r    162  until the magnetizing inductance L m    138  is fully discharged. 
     A third waveform of  FIG. 13  shows the voltage, V IN , of the DC input voltage source  180  and the voltages, V in;1  and V in;2 , of the input capacitors  122 - 1 ,  122 - 2 . It can be seen that when two primary windings are used, the voltage of the two input capacitors  122 - 1 ,  122 - 2  is equal to the half of the input voltage. 
     A fourth waveform of  FIG. 13  shows the current, i Lm , of the magnetizing inductance  138 . Notably, as depicted in  FIG. 13 , this current is discontinuous. Indeed, the current, i Lm , of the magnetizing inductance  138  is illustrated as ranging from a baseline  1302  of 0 A to an i Lm  peak  1304  of 1.5 A. 
     A fifth waveform of  FIG. 13  shows the current, i L , of the output inductor  146 . Notably, as depicted in  FIG. 13 , this current is continuous. Indeed, the current, i L , of the output inductor  146  is illustrated as ranging from a baseline  1306  of 6 A to an i L , peak  1308  of 9 A. Thus, the simulation waveforms of  FIG. 13  represent the discontinuous conduction mode of  FIG. 11 , where only one of the inductor currents is discontinuous. A sixth and seventh waveform of  FIG. 13  show currents i D1  and i D2 , which represent respective currents of the first diode  142  and the second diode  144 . An eight waveform of  FIG. 13  shows current i Dr , which represents a current through the relaxation diode D r    162 . A ninth waveform of  FIG. 13  shows V out . 
       FIG. 14  is a graph with an exemplary curve showing blocking voltage of the primary side switching devices  160  as a function of the number, N, of primary windings  132 , in accordance with some embodiments. With reference to  FIG. 14 , it can be seen that the blocking voltage of the switches reduces as N, the number of primary windings  132 , increases. The graph illustrated in  FIG. 14  has been computed based on a scenario wherein the forward converter  500  of  FIG. 5  is used for a step-down conversion ratio with the following parameters: V IN =400 V; V out =20 V; and 
     
       
         
           
             
               
                 n 
                 
                   p 
                   ; 
                   k 
                 
               
               
                 n 
                 r 
               
             
             = 
             
               
                 1 
                 4 
               
               . 
             
           
         
       
     
     In the circuit illustrated in  FIG. 5 , a blocking voltage, V ds;k , of the primary side switching device  160 - k  is given with the equation: 
     
       
         
           
             
               
                 
                   
                     V 
                     
                       
                         d 
                         ⁢ 
                         s 
                       
                       ; 
                       k 
                     
                   
                   = 
                   
                     
                       
                         V 
                         IN 
                       
                       N 
                     
                     + 
                     
                       
                         V 
                         IN 
                       
                       ⁢ 
                       
                         
                           
                             n 
                             
                               p 
                               ; 
                               k 
                             
                           
                           
                             n 
                             r 
                           
                         
                         . 
                       
                     
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
           
         
       
     
     In equation (1), V IN  is the voltage of the DC input voltage source  180 , N is the number of primary windings  132 , i.e., the number of input capacitors  122 , n p;k  represents the number of turns of the k th  primary winding  132 - k  (in one aspect of the present application, this value is equal for all primary windings, that is, n p;1 =n p;2 = . . . =n p;N ), n r  represents the number of turns of the relaxation winding  136 . 
       FIG. 15  is a graph with an exemplary curve showing blocking voltage of the primary side switches as a function of the number, N, of primary windings  132 , in accordance with some embodiments. With reference to  FIG. 15 , it can be seen that the blocking voltage of the switches reduces as N, the number of primary windings  132 , increases. The graph illustrated in  FIG. 15  has been computed based on a scenario wherein the forward converter  600  of  FIG. 6  is used for a step-down conversion ratio with the following parameters: V IN =400 V; V IN =20 V; and 
     
       
         
           
             
               
                 n 
                 
                   p 
                   ; 
                   k 
                 
               
               
                 n 
                 r 
               
             
             = 
             2. 
           
         
       
     
     In the circuit illustrated in  FIG. 6 , a blocking voltage, V ds;k , of the primary side switching devices  160  is given with the equation: 
     
       
         
           
             
               
                 
                   
                     V 
                     
                       
                         d 
                         ⁢ 
                         s 
                       
                       ; 
                       k 
                     
                   
                   = 
                   
                     
                       
                         V 
                         IN 
                       
                       N 
                     
                     + 
                     
                       
                         V 
                         out 
                       
                       ⁢ 
                       
                         
                           
                             n 
                             
                               p 
                               ; 
                               k 
                             
                           
                           
                             n 
                             r 
                           
                         
                         . 
                       
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
     In equation (2), V out  is the value of the output voltage. 
       FIG. 16  illustrates a forward converter  1600  which is similar to the forward converter  200  of  FIG. 2 , in accordance with some embodiments. That is, the forward converter  1600  includes the generic input voltage source  110 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  1600  includes a current-sensing resistor  210  connected between the generic input voltage source  110  and the capacitive divider  120 . A voltage amplifier circuit  220  is configured to measure a voltage difference across the current-sensing resistor  210  and to provide an indication of the current to the controller circuit  170 . The indication of the current provided to the controller circuit  170  may be considered representative of current through any of the primary side switching devices  160 . Advantageously, even though a single current-sensing resistor  210  is used, current through any of the primary side switching devices  160  may be measured. Thus, in accordance with the disclosed approach, one current-sensing resistor per primary side switching device  160  is not needed. Additionally, current associated with a body diode current of a primary side switching device  160  is also detected through the current-sensing resistor  210 . 
     The controller circuit  170  may base control (through control signal c 1 , c 2 , . . . , C N ) of the primary side switching devices  160 / 161  on receipt, from the output voltage conditioner  152 , of a conditioned indication of a voltage across the output capacitor  148  and receipt, from the voltage amplifier circuit  220 , of an indication of the current measured through the current-sensing resistor  210 . 
       FIG. 17  illustrates a forward converter  1700  which is similar to the forward converter  100  of  FIG. 1 , in accordance with some embodiments. That is, the forward converter  1700  includes the generic input voltage source  110 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  1700  includes a current sensing device, with its implementation shown here with the current-sensing resistor  210 , similar to that of  FIG. 16 , connected between the generic input voltage source  110  and the capacitive divider  120 . The voltage amplifier circuit  220  is arranged to measure a voltage difference across the current-sensing resistor  210  and provide an indication of the current to the controller circuit  170 . The indication of the current provided to the controller circuit  170  may be considered representative of current through any of the primary side switching devices  160  and the current through the relaxation winding  136 . The controller circuit  170  may base control (through control signal c 1 , c 2 , . . . , C N ) of the primary side switching devices  160 / 161  on receipt, from the output voltage conditioner  152 , of a conditioned indication of a voltage across the output capacitor  148  and receipt, from the voltage amplifier circuit  220 , of an indication of the current measured through the current-sensing resistor  210 . 
     Forward converters illustrated in  FIGS. 18-23  include a voltage snubbing circuit associated with a corresponding primary winding, in accordance with some embodiments. Conveniently, the voltage snubbing circuit may be adapted to reduce a voltage overshoot across the corresponding primary side switching device and discharge a transformer leakage inductance. 
       FIG. 18  illustrates a forward converter  1800  similar to the forward converter  500  of  FIG. 5 , in accordance with some embodiments. That is, the forward converter  1800  includes the DC input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the diodes  142 ,  144 , the primary side switching devices  160 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  1800  includes a passive snubber circuit connected across the N th  primary winding  132 -N. The passive snubber circuit includes a snubber diode  606  (D sn ) in series with a parallel combination of a snubber capacitor  604  (C sn ) and a snubber resistor  602  (R sn ). 
       FIG. 19  illustrates a forward converter  1900  similar to the forward converter  600  of  FIG. 6 , in accordance with some embodiments. That is, the forward converter  1900  includes the DC input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the diodes  142 ,  144 , the primary side switching devices  160 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  1900  includes the passive snubber circuit, familiar from  FIG. 18 , connected across the N th  primary winding  132 -N. 
       FIG. 20  illustrates a forward converter  2000  similar to the forward converter  500  of  FIG. 5 , in accordance with some embodiments. That is, the forward converter  2000  includes the DC input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  2000  includes a passive snubber circuit connected across the N th  primary winding  132 -N. The passive snubber circuit includes the snubber diode  606  in series with a parallel combination of the snubber capacitor  604  and a snubber Zener diode  608  (DZ sn ). 
       FIG. 21  illustrates a forward converter  2100  similar to the forward converter  600  of  FIG. 6 , in accordance with some embodiments. That is, the forward converter  2100  includes the DC input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  2100  includes the passive snubber circuit, familiar from  FIG. 20 , connected across the N th  primary winding  132 -N. 
       FIG. 22  illustrates a forward converter  2200  similar to the forward converter  500  of  FIG. 5 , in accordance with some embodiments. That is, the forward converter  2200  includes the generic input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  2200  includes an active snubber circuit connected across the N th  primary winding  132 -N. The active snubber circuit includes a series combination of the snubber capacitor  604  and an active snubber switch  610 . 
       FIG. 23  illustrates a forward converter  2300  similar to the forward converter  600  of  FIG. 6 , in accordance with some embodiments. That is, the forward converter  2100  includes the DC input voltage source  180 , the capacitive divider  120  with capacitive devices  122 , the transformer  130  with primary windings  132  and secondary winding  134 , the primary side switching devices  160 , the diodes  142 ,  144 , the output inductor  146 , the output capacitor  148 , the output load  150 , the output voltage conditioner  152 , the controller circuit  170 , the relaxation winding  136  and the relaxation diode  162 . Additionally, the forward converter  2300  includes the active snubber circuit, familiar from  FIG. 22 , connected across the N th  primary winding  132 -N. 
     The above-described implementations of the present application are intended to be examples only. Alterations, modifications and variations may be effected to the particular implementations by those skilled in the art without departing from the scope of the application, which is defined by the claims appended hereto.