Patent Publication Number: US-11653152-B1

Title: Microchip for driving a resonant circuit

Description:
The present application is a continuation of U.S. patent application Ser. No. 17/552,275 filed Dec. 15, 2021 all of which is incorporated by reference herein in its entirety. 
    
    
     FIELD 
     The present invention relates to a microchip for driving a resonant circuit. The present invention more particularly relates a microchip for driving a resonant circuit in the form of an inductance (L) capacitance (C) circuit (LC tank), an antenna or a piezoelectric transducer. 
     BACKGROUND 
     Therapeutic aerosol delivery is the mainstay for the treatment of asthma, chronic obstructive pulmonary disease (COPD) and cystic fibrosis. Therapeutic aerosol also has applications for the treatment of influenza, osteoporosis as well as the delivery of vaccines. 
     Pulmonary delivery of therapeutics for the treatment of non-respiratory systemic disease is appealing because of high lung vascularity, a thin blood-alveolar barrier, large surface area, avoidance of gastric enzymes and first-pass hepatic metabolism. It is also appealing because of improved patient comfort and adherence. The pulmonary system can be leveraged to deliver antibodies, proteins, pain killers and nucleic acids. The treatment of central nervous system disorders, such as tobacco dependence, could be significantly enhanced through the efficient delivery of nicotine to the systematic circulation through the lungs. 
     The effectiveness of therapeutic aerosol relates to the amount of drug deposited beyond the oropharyngeal region. The region where the deposit occurs is a function of the inhaled particles size. 
     The devices currently used for the administration of inhaled drugs are divided into three categories: nebulizers, metered-dose inhalers, and dry powder inhalers. Nebulizers are typically divided into two types: jet and ultrasonic but in conventional devices both types have weaknesses and present issues. 
     Jet nebulizers are based on the Bernoulli principle and produce relatively large droplets that generally deposit in the oropharyngeal region and are therefore not particularly effective. Ultrasonic nebulizers use piezoelectric crystals that vibrate at frequencies, ranging between 1 MHz and 1.7 MHz, transmitting the vibratory energy to the liquid converting it to aerosol. It is acknowledged that ultrasonic nebulizers are not effective if viscous suspensions or solutions are used and tend to heat the medication, hence destroy the molecules and remove the benefits of inhalation. 
     There are other applications which require a resonant circuit to be driven efficiently in an optimal manner at or near the resonant frequency of the resonant circuit. For instance, a device which incorporates a resonant circuit in the form of an antenna typically needs to drive the antenna with a precise AC drive signal to enable the antenna to function optimally. In addition, devices which incorporate a resonant circuit in the form an ultrasonic piezoelectric transducer must produce an AC drive signal to drive the ultrasonic transducer optimally. 
     Thus, a need exists in the art for a microchip for driving a resonant circuit, such as an LC tank, an antenna or a piezoelectric transducer, which seeks to address at least some of the problems described herein. 
     BRIEF DESCRIPTION OF THE INVENTION 
     The present invention provides a microchip as claimed in claim  1  or claim  13  and an apparatus as claimed in claim  16 . The present invention also provides preferred embodiments as claimed in the dependent claims. 
     The various examples of this disclosure which are described below have multiple benefits and advantages over conventional microchips. These benefits and advantages are set out in the description below. 
     Since the microchips and the apparatus of examples of this disclosure enable higher efficiency operation than conventional microchips and apparatuses, the microchips and apparatuses of examples of this disclosure have an environmental benefit due to the reduced power requirement. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
       In order that the present disclosure may be more readily understood, preferable embodiments thereof will now be described, by way of example only, with reference to the accompanying drawings, in which: 
         FIG.  1    is a schematic diagram of an integrated circuit arrangement of this disclosure; 
         FIG.  2    is a schematic diagram of an integrated circuit of this disclosure; 
         FIG.  3    is a schematic diagram of a pulse width modulation generator of this disclosure; 
         FIG.  4    is timing diagram of an example of this disclosure; 
         FIG.  5    is timing diagram of an example of this disclosure; 
         FIG.  6    is a table showing port functions of an example of this disclosure; 
         FIG.  7    is a schematic diagram of an integrated circuit of this disclosure; 
         FIG.  8    is a circuit diagram of an H-bridge of an example of this disclosure; 
         FIG.  9    is a circuit diagram of a current sense arrangement of an example of this disclosure; 
         FIG.  10    is a circuit diagram of an H-bridge of an example of this disclosure; 
         FIG.  11    is a graph showing the voltages during the phases of operation of the H-bridge of  FIG.  8   ; 
         FIG.  12    is a graph showing the voltages during the phases of operation of the H-bridge of  FIG.  8   ; 
         FIG.  13    is a graph showing the voltage and current at a terminal of an ultrasonic transducer while the ultrasonic transducer is being driven by the H-bridge of  FIG.  8   ; 
         FIG.  14    is a schematic diagram showing connections between integrated circuits of this disclosure; 
         FIG.  15    is a schematic diagram of an integrated circuit of this disclosure; and 
         FIG.  16    is diagram illustrating the steps of an authentication method of an example of this disclosure. 
     
    
    
     DETAILED DESCRIPTION OF THE DISCLOSURE 
     Aspects of the present disclosure are best understood from the following detailed description when read with the accompanying figures. It is noted that, in accordance with the standard practice in the industry, various features are not drawn to scale. In fact, the dimensions of the various features may be arbitrarily increased or reduced for clarity of discussion. 
     The following disclosure provides many different embodiments, or examples, for implementing different features of the provided subject matter. Specific examples of components, concentrations, applications and arrangements are described below to simplify the present disclosure. These are, of course, merely examples and are not intended to be limiting. For example, the attachment of a first feature and a second feature in the description that follows may include embodiments in which the first feature and the second feature are attached in direct contact, and may also include embodiments in which additional features may be positioned between the first feature and the second feature, such that the first feature and the second feature may not be in direct contact. In addition, the present disclosure may repeat reference numerals and/or letters in the various examples. This repetition is for the purpose of simplicity and clarity and does not in itself dictate a relationship between the various embodiments and/or configurations discussed. 
     The following disclosure describes representative examples. Each example may be considered to be an embodiment and any reference to an “example” may be changed to “embodiment” in the present disclosure. 
     Referring now to  FIG.  1    of the accompanying drawings, a driver device  202  comprises a microchip which is referred to herein as a power management integrated circuit or PMIC  300 . The PMIC  300  is a microchip for driving a resonant circuit. The resonant circuit is an LC tank, an antenna or a piezoelectric transducer. In this example, the resonant circuit is an ultrasonic transducer  215  which is provided within a resonant circuit device  201 . In this example, the resonant circuit device  201  is a separate device which is releasably coupled to the driver device  202 . In other examples, the elements of the resonant circuit device  201  are combined in the same device as the driver device  202 . 
     In this disclosure, the terms chip, microchip and integrated circuit are interchangeable. The microchip or integrated circuit is a single unit which comprises a plurality of interconnected embedded components and subsystems. The microchip is, for example, at least partly of a semiconductor, such as silicon, and is fabricated using semiconductor manufacturing techniques. 
     This disclosure describes an example resonant circuit device  201  which is a mist inhaler device which comprises an ultrasonic transducer  215 . When the ultrasonic transducer  215  is activated by the driver device  202 , the ultrasonic transducer  215  atomises a liquid to generate a mist for inhalation by a user. 
     However, it is to be appreciated that the elements of the driver device are used differently in other applications involving a resonant circuit. In these other examples, the ultrasonic transducer  215  is replaced with another resonant circuit, such as LC tank or an antenna. 
     In one example, the resonant circuit device  201  comprises a resonant circuit in the form of a piezoelectric ultrasonic transducer which generates ultrasonic waves that are used for wireless power transfer. In this example, the ultrasonic transducer is driven by the driver device  202  to generate ultrasonic waves that can be focused and sent to a receiver transducer. The receiver transducer converts the ultrasonic waves back into electrical energy and stores the energy in an energy storage device, such as a battery, or uses the electrical energy to power a device. In this way, a device can be remotely charged or powered via the power which is transferred wirelessly via the ultrasonic waves without the device having to be tethered to an electrical outlet. 
     As described below, the driver device  202  is configured to modulate the frequency and duty cycle of the AC power signal driving an ultrasonic transducer with high accuracy and efficiency. In the case of a power transfer system, this enables the driver device  202  to encode information for wireless transmission by modulating the ultrasonic waves which carry the encoded information. In one example, the driver device  202  is configured for use in a wireless power transfer system such as but not limited to the type described in U.S. Pat. No. 9,001,622 entitled “receiver communications for wireless power transfer” which is incorporated herein by reference in its entirety. 
     In another example, the driver device  202  drives a resonant circuit in the form of an ultrasonic transducer for delivering ultrasonic waves at a precise frequency and at high intensity to treat tumours. The high-intensity focused ultrasound from the ultrasonic transducer is used as a non-invasive targeted treatment to raise the temperature within a tumour to above 65° C., killing the cells of the tumour without damaging the surrounding tissue. 
     In a further application, the driver device  202  is used to drive a resonant circuit in the form an antenna to control the frequency and power of waves transmitted by the antenna accurately. In this example, the antenna may be used for any purpose. In one example, the driver device  202  drives an antenna to transmit waves at a precise frequency for the purpose of searching for materials, such as minerals or gold in the ground. 
     The driver device  202  comprises a second microchip which is referred to herein as a bridge integrated circuit or bridge IC  301  which is electrically connected to the PMIC  300 . 
     The bridge IC  301  is a microchip for driving a resonant circuit, such as an LC tank, an antenna or a piezoelectric transducer. The bridge IC  301  is a single unit which comprises a plurality of interconnected embedded components and subsystems. 
     In this example, the PMIC  300  and the bridge IC  301  are mounted to the same PCB of the driver device  202 . In this example, the physical dimensions of the PMIC  300  are 1-3 mm wide and 1-3 mm long and the physical dimensions of the bridge IC  301  are 1-3 mm wide and 1-3 mm long. 
     In this example, the resonant circuit device  201  comprises an optional programmable or one time programmable integrated circuit or OTP IC  242 . When the resonant circuit device  201  is coupled to the driver device  202 , the OTP IC is electrically connected to the PMIC  300  to receive power from the PMIC  300  such that the PMIC  300  can manage the voltage supplied to the OTP IC  242 . The OTP IC  242  is also connected to a communication bus  302  in the driver device  202 . In this example, the communication bus  302  is an I2C bus but in other examples the communication bus  302  is another type of digital serial communication bus. 
     The OTP IC  242  provides a security function which is described below. However, it is to be appreciated that the OTP IC  242  is omitted in examples of this disclosure which do not require such a security function. 
     The ultrasonic transducer  215  in the resonant circuit device  201  is electrically connected to the bridge IC  301  so that the ultrasonic transducer  215  may be driven by an AC drive signal generated by the bridge IC  301  when the device  200  is in use. 
     The driver device  202  comprises a processor in the form of a microcontroller  303  which is electrically coupled for communication with the communication bus  302 . In this example, the microcontroller  303  is a Bluetooth™ low energy (BLE) microcontroller but in other examples the microcontroller  303  is a general purpose processor. The microcontroller  303  receives power from a low dropout regulator (LDO)  304  which is driven by the battery  250 . The LDO  304  provides a stable regulated voltage to the microcontroller  303  to enable the microcontroller  303  to operate consistently even when there is a variation in the voltage of the battery  250 . 
     The driver device  202  comprises a voltage regulator in the form a DC-DC boost converter  305  which is powered by the battery  250 . The boost converter  305  increases the voltage of the battery  250  to a programmable voltage VBOOST. The programmable voltage VBOOST is set by the boost converter  305  in response to a voltage control signal VCTL from the PMIC  300 , As will be described in more detail below, the boost converter  305  outputs the voltage VBOOST to the bridge IC  301 . In other examples, the voltage regulator is a buck converter or another type of voltage regulator which outputs a selectable voltage. 
     The voltage control signal VCTL is generated by a digital to analogue converter (DAC) which, in this example, is implemented within the PMIC  300 . The DAC is not visible in  FIG.  1    since the DAC is integrated within the PMIC  300 . The DAC and the technical benefits of integrating the DAC within the PMIC  300  are described in detail below. 
     In this example, the PMIC  300  is connected to a power source connector in the form of a universal serial bus (USB) connector  306  so that the PMIC  300  can receive a charging voltage VCHRG when the USB connector  306  is coupled to a USB charger. 
     In this example, the driver device  202  comprises a first pressure sensor  307  which, in this example, is a static pressure sensor. The driver device  202  also comprises a second pressure sensor  308  which, in this example, is a dynamic pressure sensor. However, in other examples, the driver device  202  comprises only one of the two pressure sensors  307 ,  308  or the pressure sensors  307 ,  308  are omitted entirely. In this example, which relates to a mist generator device, the pressure sensors  307 ,  308  sense a change in the pressure in an aerosol chamber (not shown) to sense when a user is drawing mist from the aerosol chamber. 
     In this example, the driver device  202  comprises a plurality of LEDs  321 - 326  which are controlled by the PMIC  300 . In other examples, the LEDs  321 - 326  are omitted. 
     The microcontroller  303  functions as a master device on the communication bus  302 , with the PMIC  300  being a first slave device, the OTP IC  242  being a second slave device, the second pressure sensor  308  being a third slave device and the first pressure sensor  307  being the a fourth slave device. The communication bus  302  enables the microcontroller  303  to control the following functions within the driver device  202 :
         1. All functions of the PMIC are highly configurable by the microcontroller  303 .   2. The current flowing through the resonant circuit (ultrasonic transducer  215 ) is sensed by a high bandwidth sense and rectifier circuit at a high common mode voltage (high side of the bridge). The sensed current is converted into a voltage proportional to the rms current and provided as a buffered voltage at a current sense output pin  309  of the bridge IC  301 . This voltage is fed to and sampled in the PMIC  300  and made available as a digital representation via I2C requests. Sensing the current flowing through the ultrasonic transducer  215  forms part of the resonant frequency tracking functionality. As described herein, the ability of the device to enable this functionality within the bridge IC  301  provides significant technical benefits.   3. The DAC (not shown in  FIG.  1   ) integrated within the PMIC  300  enables the DC-DC boost converter voltage VBOOST to be programmed to be between 10V and 20V.   4. The microcontroller  303  enables the charger sub-system of the driver device  202  to manage the charging of the battery  250 , which in this example is a single cell battery.   5. A Light Emitting Diode (LED) driver module (not shown) is powered by the PMIC  300  to drive and dim digitally the LEDs  321 - 326  either in linear mode or in gamma corrected mode.   6. The microcontroller  303  is able to read Pressure #1 and Pressure #2 sensor values from the pressure sensors  307 ,  308 .       

     Referring now to  FIG.  2    of the accompanying drawings, the PMIC  300  is, in this example, a self-contained chip or integrated circuit which comprises integrated subsystems and a plurality of pins which provide electrical inputs and outputs to the PMIC  300 . The references to an integrated circuit or chip in this disclosure are interchangeable and either term encompasses a semiconductor device which may, for instance, be of silicon. 
     The PMIC  300  comprises an analogue core  310  which comprises analogue components including a reference block (BG)  311 , a LDO  312 , a current sensor  313 , a temperature sensor  314  and an oscillator  315 . 
     As described in more detail below, the oscillator  315  is coupled to a delay locked loop (DLL) which outputs pulse width modulation (PWM) phases A and B. The oscillator  315  and the DLL generate a two phase centre aligned PWM output which drives an H bridge in the bridge IC  301 . 
     The DLL comprises a plurality of delay lines connected end to end, wherein the total delay of the delay lines is equal to the period of the main clock signal clk_m. In this example, the DLL is implemented in a digital processor subsystem, referred to herein as a digital core  316 , of the PMIC  300  which receives a clock signal from the oscillator  315  and a regulated power supply voltage from the LDO  312 . The DLL is implemented in a large number (e.g. in the order of millions) of delay gates which are connected end to end in the digital core  316 . 
     The implementation of the oscillator  315  and the DLL in the same integrated circuit of the PMIC  300  in order to generate a two phase centre aligned PWM signal is unique since at present no signal generator component in the integrated circuit market comprises this implementation. 
     As described herein, PWM is part of the functionality which enables the driver device  202  to track the resonant frequency of the ultrasonic transducer  215  accurately in order to maintain an efficient transfer from electrical energy to kinetic energy in order to optimise the generation of mist. The same functionality enables other examples which comprise a different resonant circuit by driving the resonant circuit efficiently and at a high power and a high frequency. 
     In this example, the PMIC  300  comprises a charger circuit  317  which controls the charging of the battery  250 , for instance by power from a USB power source. The charger circuit  317  is omitted in other examples which do not require a battery. 
     The PMIC  300  comprises an integrated power switch VSYS which configures the PMIC  300  to power the analogue core  310  by power from the battery  250  or by power from an external power source if the battery  250  is being charged. 
     The PMIC  300  comprises an embedded analogue to digital converter (ADC) subsystem  318 . The implementation of the ADC  318  together with the oscillator  315  in the same integrated circuit is, in itself, unique since there is no other integrated circuit in the integrated circuit market which comprises an oscillator and an ADC implemented as sub-blocks within the integrated circuit. In a conventional device, an ADC is typically provided as a separate discrete component from an oscillator with the separate ADC and oscillator being mounted to the same PCB. The problem with this conventional arrangement is that the two separate components of the ADC and the oscillator take up space unnecessarily on the PCB. A further problem is that the conventional ADC and oscillator are usually connected to one another by a serial data communication bus, such as an I2C bus, which has a limited communication speed of up to only 400 kHz. In contrast to conventional devices, the PMIC  300  comprises the ADC  318  and the oscillator  315  integrated within the same integrated circuit which eliminates any lag in communication between the ADC  318  and the oscillator  315 , meaning that the ADC  318  and the oscillator  315  can communicate with one another at high speed, such as at the speed of the oscillator  315  (e.g. 3 MHz to 5 MHz). 
     In the PMIC  300  of this example, the oscillator  315  is running at 5 MHz and generates a clock signal SYS CLOCK at 5 MHz. However, in other examples, the oscillator  315  generates a clock signal at a much higher frequency of up to 105 MHz. The integrated circuits described herein are all configured to operate at the high frequency of the oscillator  315 . 
     The ADC  318  comprises a plurality of feedback input terminals or analogue inputs  319  which comprise a plurality of GPIO inputs (IF_GPIO1-3). At least one of the feedback input terminals or the analogue inputs  319  receives a feedback signal from an H-bridge circuit in the bridge IC  301 , the feedback signal being indicative of a parameter of the operation of the H-bridge circuit or an AC drive signal when the H-bridge circuit is driving a resonant circuit, such as the ultrasonic transducer  215 , with the AC drive signal. As described below, the GPIO inputs are used to receive a current sense signal from the bridge IC  301  which is indicative of the route mean square (rms) current reported by the bridge IC  301 . In this example, one of the GPIO inputs is a feedback input terminal which receives a feedback signal from the H-bridge in the bridge IC  301 . 
     The ADC subsystem  318  samples analogue signals received at the plurality of ADC input terminals  319  at a sampling frequency which is proportional to the frequency of the main clock signal. The ADC subsystem  318  then generates ADC digital signals using the sampled analogue signals. 
     In this example, the ADC  318  which is incorporated in the PMIC  300  samples not only the RMS current flowing through the H-bridge  334  and the ultrasonic transducer  215  but also voltages available in the system (e.g. VBAT, VCHRG, VBOOST), the temperature of the PMIC  300 , the temperature of the battery  250  and the GPIO inputs (IF_GPIO1-3) which allow for future extensions. 
     The digital core  316  receives the ADC generated digital signals from the ADC subsystem and processes the ADC digital signals to generate the driver control signal. The digital core  316  communicates the driver control signal to the PWM signal generator subsystem (DLL  332 ) to control the PWM signal generator subsystem. 
     Rectification circuits existing in the market today have a very limited bandwidth (typically less than 1 MHz). Since the oscillator  315  of the PMIC  300  is running at up to 5 MHz or even up to 105 Mhz, a high bandwidth rectifier circuit is implemented in the PMIC  300 . As will be described below, sensing the RMS current within an H bridge of the bridge IC  301  forms part of a feedback loop which enables the driver device  202  to drive the ultrasonic transducer  215  with high precision. The feedback loop is a game changer in the industry of driving ultrasound transducers since it accommodates for any process variation in the piezo electric transducer production (variations of resonance frequencies) and it compensates for temperature effects of the resonance frequency. This is achieved, in part, by the inventive realisation of integrating the ADC  318 , the oscillator  315  and the DLL within the same integrated circuit of the PMIC  300 . The integration enables these sub-systems to communicate with one another at high speed (e.g. at the clock frequency of 5 MHz or up to 105 MHz). Reducing the lag between these subsystems is a game changer in the ultrasonics industry, particularly in the field of mist generator devices. 
     The ADC  318  comprises a battery voltage monitoring input VBAT and a charger input voltage monitoring input VCHG as well as voltage monitoring inputs VMON and VRTH as well as a temperature monitoring input TEMP. 
     The temperature monitoring input TEMP receives a temperature signal from the temperature sensor  314  which is embedded within the PMIC  300 . This enables the PMIC  300  to sense the actual temperature within the PMIC  300  accurately so that the PMIC  300  can detect any malfunction within the PMIC  300  as well as malfunction to other components on the printed circuit board which affect the temperature of the PMIC  300 . The PMIC  300  can then control the bridge IC  301  to prevent excitation of the ultrasonic transducer  215  if there is a malfunction in order to maintain the safety of the resonant circuit device  201 . 
     The additional temperature sensor input VRTH receives a temperature sensing signal from an external temperature sensor within the driver device  202  which monitors the temperature of a battery. The PMIC  300  can thus react to stop the battery from being charged in the event of a high battery temperature or otherwise shut down the driver device  202  in order to reduce the risk of damage being caused by an excessively high battery temperature. 
     The PMIC  300  comprises an LED driver  320  which, in this example, receives a digital drive signal from the digital core  316  and provides LED drive output signals to six LEDs  321 - 326  which are configured to be coupled to output pins of the PMIC  300 . The LED driver  320  can thus drive and dim the LEDs  321 - 326  in up to six independent channels. 
     The PMIC  300  comprises a first digital to analogue converter (DAC)  327  which converts digital signals within the PMIC  300  into an analogue voltage control signal which is output from the PMIC  300  via an output pin VDAC0. The first DAC  327  converts a digital control signal generated by the digital core  316  into an analogue voltage control signal which is output via the output pin VDAC0 to control a voltage regulator circuit, such as the boost converter  305 . The voltage control signal thus controls the voltage regulator circuit to generate a predetermined voltage for modulation by the H-bridge circuit to drive a resonant circuit, such as the ultrasonic transducer  215 , in response to feedback signals which are indicative of the operation of the resonant circuit (the ultrasonic transducer  215 ). 
     In this example, the PMIC  300  comprises a second DAC  328  which converts digital signals within the PMIC  300  into an analogue signal which is output from the PMIC  300  via a second analogue output pin VDAC1. 
     Embedding the DAC  327  or the DACs  327 ,  328  within the same microchip as the other subsystems of the PMIC  300  allows the DACs  327 ,  328  to communicate with the digital core  316  and other components within the PMIC  300  at high speed with no or minimal communication lag. The DACs  327 ,  328  provide analogue outputs which control external feedback loops. For instance, the first DAC  327  provides the control signal VCTL to the boost converter  305  to control the operation of the boost converter  305 . In other examples, the DACs  327 ,  328  are configured to provide a drive signal to a DC-DC buck converter instead of or in addition to the boost converter  305 . Integrating the two independent DAC channels in the PMIC  300  enables the PMIC  300  to manipulate the feedback loop of any regulator used in the driver device  202  and allows the driver device  202  to regulate the sonication power of the ultrasonic transducer  215  or to set analogue thresholds for absolute maximum current and temperature settings of the ultrasonic transducer  215 . 
     The PMIC  300  comprises a serial communication interface which, in this example, is an I2C interface which incorporates external I2C address set through pins. 
     The PMIC  300  also comprises various functional blocks which include a digital machine (FSM) to implement the functionality of the microchip. These blocks will be described in more detail below. 
     Referring now to  FIG.  3    of the accompanying drawings, a pulse width modulation (PWM) signal generator subsystem  329  is embedded within the PMIC  300 . The PWM generator system  329  comprises the oscillator  315 , and frequency divider  330 , a multiplexer  331  and a delay locked loop (DLL)  332 . As will be described below, the PWM generator system  329  is a two phase centre aligned PWM generator. 
     The frequency divider  330 , the multiplexer  331  and the DLL  332  are implemented in digital logic components (e.g. transistors, logic gates, etc.) within the digital core  316 . 
     In examples of this disclosure, the frequency range which is covered by the oscillator  315  and respectively by the PWM generator system  329  is 50 kHz to 5 MHz or up to 105 MHz. The frequency accuracy of the PWM generator system  329  is ±1% and the spread over temperature is ±1%. In the IC market today, no IC has an embedded oscillator and two phase centre aligned PWM generator that can provide a frequency range of 50 kHz to 5 MHz or up to 105 MHz. 
     The oscillator  315  generates a main clock signal (clk_m) with a frequency of 50 kHz to 5 MHz or up to 105 MHz. The main clock clk_m is input to the frequency divider  330  which divides the frequency of the main clock clk_m by one or more predetermined divisor amounts. In this example, the frequency divider  330  divides the frequency of the main clock clk_m by 2, 4, 8 and 16 and provides the divided frequency clocks as outputs to the multiplexer  331 . The multiplexer  331  multiplexes the divided frequency clocks and provides a divided frequency output to the DLL  332 . This signal which is passed to the DLL  332  is a frequency reference signal which controls the DLL  332  to output signals at a desired frequency. In other examples, the frequency divider  330  and the multiplexer  331  are omitted. 
     The oscillator  315  also generates two phases; a first phase clock signal Phase 1 and a second phase clock signal Phase 2, The phases of the first phase clock signal and the second phase clock signal are centre aligned. As illustrated in  FIG.  4   :
         The first phase clock signal Phase 1 is high for a variable time of clk_m&#39;s positive half-period and low during clk_m&#39;s negative half-period.   The second phase clock signal Phase 2 is high for a variable time of clk_m&#39;s negative half-period and low during clk_m&#39;s positive half-period.       

     Phase 1 and Phase 2 are then sent to the DLL  332  which generates a double frequency clock signal using the first phase clock signal Phase 1 and the second phase clock signal Phase 2. The double frequency clock signal is double the frequency of the main clock signal clk_m. In this example, an “OR” gate within the DLL  332  generates the double frequency clock signal using the first phase clock signal Phase 1 and the second phase clock signal Phase 2. This double frequency clock or the divided frequency coming from the frequency divider  330  is selected based on a target frequency selected and then used as reference for the DLL  332 . 
     Within the DLL  332 , a signal referred to hereafter as “clock” represents the main clock clk_m multiplied by 2, while a signal referred to hereafter as “clock_del” is a replica of clock delayed by one period of the frequency. Clock and clock_del are passed through a phase frequency detector. A node Vc is then charged or discharged by a charge-pump based on the phase error polarity. A control voltage is fed directly to control the delay of every single delay unit within the DLL  332  until the total delay of the DLL  332  is exactly one period. 
     The DLL  332  controls the rising edge of the first phase clock signal Phase 1 and the second phase clock signal Phase 2 to be synchronous with the rising edge of the double frequency clock signal. The DLL  332  adjusts the frequency and the duty cycle of the first phase clock signal Phase 1 and the second phase clock signal Phase 2 in response to a respective frequency reference signal and a duty cycle control signal to produce a first phase output signal Phase A and a second phase output signal Phase B to drive an H-bridge or an inverter to generate an AC drive signal to drive an ultrasonic transducer. 
     The PMIC  300  comprises a first phase output signal terminal PHASE_A which outputs the first phase output signal Phase A to an H-bridge circuit and a second phase output signal terminal PHASE_B which outputs the second phase output signal Phase B to an H-bridge circuit. 
     In this example, the DLL  332  adjusts the duty cycle of the first phase clock signal Phase 1 and the second phase clock signal Phase 2 in response to the duty cycle control signal by varying the delay of each delay line in the DLL  332  response to the duty cycle control signal. 
     The clock is used at double of its frequency because guarantees better accuracy. As shown in  FIG.  5   , for the purpose of explanation if the frequency of the main clock clk_m is used (which it is not in examples of this disclosure), Phase A is synchronous with clock&#39;s rising edge R, while Phase B is synchronous with clock&#39;s falling edge F. The delay line of the DLL  332  controls the rising edge R and so, for the falling edge F, the PWM generator system  329  would need to rely on a perfect matching of the delay units of the DLL  332  which can be imperfect. However, to remove this error, the PWM generator system  329  uses the double frequency clock so that both Phase A and Phase B are synchronous with the rising edge R of the double frequency clock. 
     To perform a duty-cycle from 20% to 50% with a 2% step size, the delay line of the DLL  332  comprises 25 delay units, with the output of each respective delay unit representing a Phase nth. Eventually the phase of the output of the final delay unit will correspond to the input clock, Considering that all delays will be almost the same, a particular duty cycle is obtained with the output of the specific delay unit with simple logic in the digital core  316 . 
     It is important to take care of the DLL  332  startup as the DLL  332  might not be able to lock a period of delay but two or more periods, taking the DLL  332  to a non-convergence zone. To avoid this issue, a start-up circuit is implemented in the PWM generator system  329  which allows the DLL  332  to start from a known and deterministic condition. The start-up circuit furthermore allows the DLL  332  to start with the minimum delay. 
     In examples of this disclosure, the frequency range covered by the PWM generator system  329  is extended and so the delay units in the DLL  332  can provide delays of 4 ns (for an oscillator frequency of 5 MHz) to 400 ns (for an oscillator frequency of 50 kHz). In order to accommodate for these differing delays, capacitors Cb are included in the PWM generator system  329 , with the capacitor value being selected to provide the required delay. 
     The Phase A and Phase B are output from the DLL  332  and passed through a digital  10  to the bridge IC  301  so that the Phase A and Phase B can be used to control the operation of the bridge IC  301 . 
     The battery charging functionality of the driver device  202  will now be described in more detail. The battery charging sub-system comprises the charger circuit  317  which is embedded in the PMIC  300  and controlled by a digital charge controller hosted in the PMIC  300 . The charger circuit  317  is controlled by the microcontroller  303  via the communication bus  302 . The battery charging sub-system is able to charge a single cell lithium polymer (LiPo) or lithium-ion (Won) battery. 
     In this example, the battery charging sub-system is able to charge a battery or batteries with a charging current of up to 1 A from a 5V power supply (e.g. a USB power supply). One or more of the following parameters can be programmed through the communication bus  302  (I2C interface) to adapt the charge parameters for the battery:
         Charge voltage can be set between 3.9V and 4.3V in 100 mV steps.   The charge current can be set between 150 mA and 1000 mA in 50 mA steps.   The pre-charge current is 1/10 of the charge current.   Pre-charge and fast charge timeouts can be set between 5 and 85 min respectively 20 and 340 min.   Optionally an external negative temperature coefficient (NTC) thermistor can be used to monitor the battery temperature.       

     In some examples, the battery charging sub-system reports one or more of the following events by raising an interrupt to the host microcontroller  303 ; Battery detected
         Battery is being charged   Battery is fully charged   Battery is not present   Charge timeout reached   Charging supply is below the undervoltage limit       

     The main advantage of having the charger circuit  317  embedded in the PMIC  300 , is that it allows all the programming options and event indications listed to be implemented within the PMIC  300  which guarantees the safe operation of the battery charging sub-system. Furthermore, a significant manufacturing cost and PCB space saving can be accomplished compared with conventional resonant circuit devices, such as mist inhaler devices, which comprise discrete components of a charging system mounted separately on a PCB. The charger circuit  317  also allows for highly versatile setting of charge current and voltage, different fault timeouts and numerous event flags for detailed status analysis. 
     The analogue to digital converter (ADC)  318  will now be described in more detail. The inventors had to overcome significant technical challenges to integrate the ADC  318  within the PMIC  300  with the high speed oscillator  315 . Moreover, integrating the ADC  318  within the PMIC  300  goes against the conventional approach in the art which relies on using one of the many discrete ADC devices that are available in the IC market. 
     In this example, the ADC  318  samples at least one parameter within the ultrasonic transducer driver chip (PMIC  300 ) at a sampling rate which is equal to the frequency of the main clock signal clk_m. In this example, the ADC  318  is a 10 bit analogue to digital converter which is able to unload digital sampling from the microprocessor  303  to save the resources of the microprocessor  303 . Integrating the ADC  318  within the PMIC  300  also avoids the need to use an I2C bus that would otherwise slow down the sampling ability of the ADC (a conventional device relies on an I2C bus to communicate data between a dedicated discrete ADC and a microcontroller at a limited clock speed of typically up to 400 kHz). 
     In examples of this disclosure, one or more of the following parameters can be sampled sequentially by the ADC  318 :
         i. An rms current signal which is received at the ultrasonic transducer driver chip (PMIC  300 ) from an external inverter circuit which is driving an ultrasonic transducer. In this is example, this parameter is a root mean square (rms) current reported by the bridge IC  301 . Sensing the rms current is important to implementing the feedback loop used for driving the ultrasound transducer  215 . The ADC  318  is able to sense the rms current directly from the bridge IC  301  via a signal with minimal or no lag since the ADC  318  does not rely on this information being transmitted via an I2C bus. This provides a significant speed and accuracy benefit over conventional devices which are constrained by the comparatively low speeds of an I2C bus.   ii. The voltage of a battery connected to the PMIC  300 .   iii. The voltage of a charger connected to the PMIC  300 .   iv. A temperature signal, such as a temperature signal which is indicative of the PMIC  300  chip temperature. As described above, this temperature can be measured very accurately due to the temperature sensor  314  being embedded in the same IC as the oscillator  315 . For example, if the PMIC  300  temperature goes up, the current, frequency and PWM are regulated by the PMIC  300  to control the transducer oscillation which in turn controls the temperature.   v. Two external pins.   vi. External NTC temperature sensor to monitor battery pack temperature.       

     In some examples, the ADC  318  samples one or more of the above-mentioned sources sequentially, for instance in a round robin scheme. The ADC  318  samples the sources at high speed, such as the speed of the oscillator  315  which may be up to 5 MHz or up to 105 MHz. 
     In some examples, the driver device  202  is configured so that a user or the manufacturer of the device can specify how many samples shall be taken from each source for averaging. For instance, a user can configure the system to take 512 samples from the rms current input, 64 samples from the battery voltage, 64 from the charger input voltage, 32 samples from the external pins and 8 from the NTC pin. Furthermore, the user can also specify if one of the above-mentioned sources shall be skipped. 
     In some examples, for each source the user can specify two digital thresholds which divide the full range into a plurality of zones, such as 3 zones. Subsequently the user can set the system to release an interrupt when the sampled value changes zones e.g. from a zone  2  to a zone  3 . 
     No conventional IC available in the market today can perform the above features of the PMIC  300 . Sampling with such flexibility and granularity is paramount when driving a resonant circuit or component, such as an ultrasound transducer. 
     In this example, the PMIC  300  comprises an 8 bit general purpose digital input output port (GPIO). Each port can be configured as digital input and digital output. Some of the ports have an analogue input function, as shown in the table in  FIG.  6   . 
     The GPIO7-GPIO5 ports of the PMIC  300  can be used to set the device&#39;s address on the communication (I2C) bus  302 . Subsequently eight identical devices can be used on the same I2C bus. This is a unique feature in the IC industry since it allows eight identical devices to be used on the same I2C bus without any conflicting addresses. This is implemented by each device reading the state of GPIO7-GPIO5 during the first 100 μs after the startup of the PMIC  300  and storing that portion of the address internally in the PMIC  300 . After the PMIC  300  has been started up the GPIOs can be used for any other purpose. 
     As described above, the PMIC  300  comprises a six channel LED driver  320 . In this example the LED driver  320  comprises N-Channel Metal-Oxide Semiconductor (NMOS) current sources which are 5V tolerant. The LED driver  320  is configured to set the LED current in four discrete levels; 5 mA, 10 mA, 15 mA and 20 mA. The LED driver  320  is configured to dim each LED channel with a 12 bit PWM signal either with or without gamma correction. The LED driver  320  is configured to vary the PWM frequency from 300 Hz to 1.5 KHz. This feature is unique in the field of resonant circuit devices, such as ultrasonic mist inhaler devices, as the functionality is embedded as a sub-system of the PMIC  300 . 
     In this example, the PMIC  300  comprises two independent 6 Bit Digital to Analog Converters (DAC)  327 ,  328  which are incorporated into the PMIC  300 . The purpose of the DACs  327 ,  328  is to output an analogue voltage to manipulate the feedback path of an external regulator (e.g. the DC-DC Boost converter  305  a Buck converter or a LDO). 
     Furthermore, in some examples, the DACs  327 ,  328  can also be used to dynamically adjust the over current shutdown level of the bridge IC  301 , as described below. 
     The output voltage of each DAC  327 ,  328  is programmable between 0V and 1.5V or between 0V and V_battery (Vbat). In this example, the control of the DAC output voltage is done via I2C commands. Having two DAC incorporated in the PMIC  300  is unique and will allow the dynamic monitoring control of the current. If either DAC  327 ,  328  was an external chip, the speed would fall under the same restrictions of speed limitations due to the I2C protocol. The active power monitoring arrangement of the driver device  202  works with optimum efficiency if all these embedded features are in the PMIC. Had they been external components, the active power monitoring arrangement would be totally inefficient. 
     Referring now to  FIG.  7    of the accompanying drawings, the bridge IC  301  is a microchip which comprises an embedded power switching circuit  333 . In this example, the power switching circuit  333  is an H-bridge  334  which is shown in  FIG.  8    and which is described in detail below. It is, however, to be appreciated that the bridge IC  301  of other examples may incorporate an alternative power switching circuit to the H-bridge  334 , provided that the power switching circuit performs an equivalent function for generating an AC drive signal to drive the ultrasonic transducer  215 . 
     The bridge IC  301  comprises a first phase terminal PHASE A which receives a first phase output signal Phase A from the PWM signal generator subsystem of the PMIC  300 . The bridge IC  301  also comprises a second phase terminal PHASE B which receives a second phase output signal Phase B from the PWM signal generator subsystem of the PMIC  300 . 
     The bridge IC  301  comprises a current sensing circuit  335  which senses current flow in the H-bridge  334  directly and provides an RMS current output signal via the RMS_CURR pin of the bridge IC  301 . The current sensing circuit  335  is configured for over current monitoring, to detect when the current flowing in the H-bridge  334  is above a predetermined threshold. The integration of the power switching circuit  333  comprising the H-bridge  334  and the current sensing circuit  335  all within the same embedded circuit of the bridge IC  301  is a unique combination in the IC market. At present, no other integrated circuit in the IC market comprises an H-bridge with embedded circuitry for sensing the RMS current flowing through the H-bridge. 
     The bridge IC  301  comprises a temperature sensor  336  which includes over temperature monitoring. The temperature sensor  336  is configured to shut down the bridge IC  301  or disable at least part of the bridge IC  336  in the event that the temperature sensor  336  detects that the bridge IC  301  is operating at a temperature above a predetermined threshold. The temperature sensor  336  therefore provides an integrated safety function which prevents damage to the bridge IC  301  or other components within the driver device  202  in the event that the bridge IC  301  operates at an excessively high temperature. 
     The bridge IC  301  comprises a digital state machine  337  which is integrally connected to the power switching circuit  333 . The digital state machine  337  receives the phase A and phase B signals from the PMIC  300  and an ENABLE signal, for instance from the microcontroller  303 . The digital state machine  337  generates timing signals based on the first phase output signal Phase A and the second phase output signal Phase B. 
     The digital state machine  337  outputs timing signals corresponding to the phase A and phase B signals as well as a BRIDGE PR and BRIDGE EN signals to the power switching circuit  333  in order to control the power switching circuit  333 . The digital state machine  337  thus outputs the timing signals to the switches T 1 -T 4  of the H-bridge circuit  334  to control the switches T 1 -T 4  to turn on and off in a sequence such that the H-bridge circuit outputs an AC drive signal for driving a resonant circuit, such as the ultrasonic transducer  215 . 
     As described in more detail below, the switching sequence comprises a free-float period in which the first switch T 1  and the second switch T 1  are turned off and the third switch T 3  and the fourth switch T 4  are turned on in order to dissipate energy stored by the resonant circuit (the ultrasonic transducer  215 ). 
     The bridge IC  301  comprises a test controller  338  which enables the bridge IC  301  to be tested to determine whether the embedded components within the bridge IC  301  are operating correctly. The test controller  338  is coupled to TEST_DATA, TEST_CLK and TEST_LOAD pins so that the bridge IC  301  can be connected to an external control device which feeds data into and out from the bridge IC  301  to test the operation of the bridge IC  301 . The bridge IC  301  also comprises a TEST BUS which enables the digital communication bus within the bridge IC  301  to be tested via a TST_PAD pin. 
     The bridge IC  301  comprises a power on reset circuit (POR)  339  which controls the startup operation of the bridge IC  301 . The POR  339  ensures that the bridge IC  301  starts up properly only if the supply voltage is within a predetermined range. If the power supply voltage is outside of the predetermined range, for instance if the power supply voltage is too high, the POR  339  delays the startup of the bridge IC  301  until the supply voltage is within the predetermined range. 
     The bridge IC  301  comprises a reference block (BG)  340  which provides a precise reference voltage for use by the other subsystems of the bridge IC  301 . 
     The bridge IC  301  comprises a current reference  341  which provides a precise current to the power switching circuit  333  and/or other subsystems within the bridge IC  301 , such as the current sensor  335 . 
     The temperature sensor  336  monitors the temperature of the silicon of the bridge IC  301  continuously. If the temperature exceeds the predetermined temperature threshold, the power switching circuit  333  is switched off automatically. In addition, the over temperature may be reported to an external host to inform the external host that an over temperature event has occurred. 
     The digital state machine (FSM)  337  generates the timing signals for the power switching circuit  333  which, in this example, are timing signals for controlling the H-bridge  334 . 
     The bridge IC  301  comprises comparators  342 , 343  which compare signals from the various subsystems of the bridge IC  301  with the voltage and current references  340 , 341  and provide reference output signals via the pins of the bridge IC  301 . 
     Referring again to  FIG.  8    of the accompanying drawings, the H-bridge  334  of this example comprises four switches in the form of NMOS field effect transistors (FET) switches on both sides of the H-bridge  334 . The H-bridge  334  comprises four switches or transistors T 1 -T 4  which are connected in an H-bridge configuration, with each transistor T 1 -T 4  being driven by a respective logic input A-D. The transistors T 1 -T 4  are configured to be driven by a bootstrap voltage which is generated internally with two external capacitors Cb which are connected as illustrated in  FIG.  8   . 
     The H-bridge  334  comprises various power inputs and outputs which are connected to the respective pins of the bridge IC  301 . The H-bridge  334  receives the programmable voltage VBOOST which is output from the boost converter  305  via a first power supply terminal, labelled VBOOST in  FIG.  8   . The H-bridge  334  comprises a second power supply terminal, labelled VSS_P in  FIG.  8   . 
     The H-bridge  334  comprises outputs OUTP, OUTN which are configured to connect to respective terminals of the ultrasonic transducer  215  so that the AC drive signal output from the H-bridge  334  can drive the ultrasonic transducer  215 . 
     The switching of the four switches or transistors T 1 -T 4  is controlled by switching signals from the digital state machine  337  via the logic input A-D. It is to be appreciated that, while  FIG.  8    shows four transistors T 1 -T 4 , in other examples, the H-bridge  334  incorporates a larger number of transistors or other switching components to implement the functionality of the H-bridge. 
     In this example, the H-bridge  334  operates at a switching power of 22 W to 50 W in order to deliver an AC drive signal with sufficient power to drive the ultrasonic transducer  215  optimally at or near the resonant frequency of the ultrasonic transducer  215 . The voltage which is switched by the H-bridge  334  of this example is ±15 V. In other examples, the voltage is ±20 V. 
     In this example, the H-bridge  334  switches at a frequency of 3 MHz to 5 MHz or up to 105 MHz. This is a high switching speed compared with conventional integrated circuit H-bridges which are available in the IC market. For instance, a conventional integrated circuit H-bridge available in the IC market today is configured to operate at a maximum frequency of only 2 MHz. Aside from the bridge IC  301  described herein, no conventional integrated circuit H-bridge available in the IC market is able to operate at a power of 22 V to 50 V at a frequency of up to 5 MHz, let alone up to 105 MHz. 
     Referring now to  FIG.  9    of the accompanying drawings, the current sensor  335  comprises positive and negative current sense resistors RshuntP, RshuntN which are connected in series with the respective high and low sides of the H-bridge  334 , as shown in  FIG.  8   . The current sense resistors RshuntP, RshuntN are low value resistors which, in this example, are 0.1Ω. The current sensor  335  comprises a first voltage sensor in the form of a first operational amplifier  344  which measures the voltage drop across the first current sensor resistor RshuntP and a second voltage sensor in the form of a second operational amplifier  345  which measures the voltage drop across the second current sensor resistor RshuntN. In this example, the gain of each operational amplifier  344 ,  345  is 2V/V. The output of each operational amplifier  344 ,  345  is, in this example, 1 mA/V. The current sensor  335  comprises a pull down resistor Rcs which, in this example, is 2 kΩ. The outputs of the operational amplifiers  344 ,  345  provide an output CSout which passes through a low pass filter  346  which removes transients in the signal CSout. An output Vout of the low pass filter  346  is the output signal of the current sensor  335 . 
     The current sensor  335  thus measures the AC current flowing through the H-bridge  334  and respectively through the ultrasonic transducer  215 , The current sensor  335  translates the AC current into an equivalent RMS output voltage (Vout) relative to ground. The current sensor  335  has high bandwidth capability since the H-bridge  334  can be operated at a frequency of up to 5 MHz or, in some examples, up to 105 MHz. The output Vout of the current sensor  335  reports a positive voltage which is equivalent to the measured AC rms current flowing through the ultrasonic transducer  215 . The output voltage Vout of the current sensor  335  is, in this example, fed back to the control circuitry within the bridge IC  301  to enable the bridge IC  301  to shut down the H-bridge  334  in the event that the current flowing through the H-bridge  334  and hence through the transducer  215  is in excess of a predetermined threshold. In addition, the over current threshold event is reported to the first comparator  342  in the bridge IC  301  so that the bridge IC  301  can report the over current event via the OVC_TRIGG pin of the bridge IC  301 . 
     Referring now to  FIG.  10    of the accompanying drawings, the control of the H-bridge  334  will now be described also with reference to the equivalent piezoelectric model of the ultrasonic transducer  215 . 
     To develop a positive voltage across the outputs OUTP, OUTN of the H-bridge  334  as indicated by V_out in  FIG.  10    (note the direction of the arrow) the switching sequence of the transistors T 1 -T 4  via the inputs A-D is as follows:
         1. Positive output voltage across the ultrasonic transducer  215 : A-ON, B-OFF, C-OFF, D-ON   2. Transition from positive output voltage to zero: A-OFF, B-OFF, C-OFF, D-ON. During this transition, C is switched off first to minimise or avoid power loss by minimising or avoiding current flowing through A and C if there is a switching error or delay in A.   3. Zero output voltage: A-OFF, B-OFF, C-ON, D-ON. During this zero output voltage phase, the terminals of the outputs OUTP, OUTN of the H-bridge  334  are grounded by the C and D switches which remain on. This dissipates the energy stored by the capacitors in the equivalent circuit of the ultrasonic transducer, which minimises the voltage overshoot in the switching waveform voltage which is applied to the ultrasonic transducer.   4. Transition from zero to negative output voltage: A-OFF, B-OFF, C-ON, D-OFF.   5. Negative output voltage across the ultrasonic transducer  215 : A-OFF, B-ON, C-ON, D-OFF       

     At high frequencies of up to 5 MHz or even up to 105 MHz, it will be appreciated that the time for each part of the switching sequence is very short and in the order of nanoseconds or picoseconds. For instance, at a switching frequency of 6 MHz, each part of the switching sequence occurs in approximately 80 ns. 
     A graph showing the output voltage OUTP, OUTN of the H-bridge  334  according to the above switching sequence is shown in  FIG.  11    of the accompanying drawings. The zero output voltage portion of the switching sequence is included to accommodate for the energy stored by the ultrasonic transducer  215  (e.g. the energy stored by the capacitors in the equivalent circuit of the ultrasonic transducer). As described above, this minimises the voltage overshoot in the switching waveform voltage which is applied to the ultrasonic transducer and hence minimises unnecessary power dissipation and heating in the ultrasonic transducer. 
     Minimising or removing voltage overshoot also reduces the risk of damage to transistors in the bridge IC  301  by preventing the transistors from being subject to voltages in excess of their rated voltage. Furthermore, the minimisation or removal of the voltage overshoot enables the bridge IC  301  to drive the ultrasonic transducer accurately in a way which minimises disruption to the current sense feedback loop described herein. Consequently, the bridge IC  301  is able to drive the ultrasonic transducer at a high power of 22 W to 50 W or even as high as 70 W at a high frequency of up to 5 MHz or even up to 105 MHz. 
     The bridge IC  301  of this example is configured to be controlled by the PMIC  300  to operate in two different modes, referred to herein as a forced mode and a native frequency mode. These two modes of operation are novel over existing bridge ICs. In particular, the native frequency mode is a major innovation which offers substantial benefits in the accuracy and efficiency of driving an ultrasonic transducer as compared with conventional devices. 
     Forced Frequency Mode (FFM) 
     In the forced frequency mode the H-bridge  334  is controlled in the sequence described above but at a user selectable frequency. As a consequence, the H-bridge transistors T 1 -T 4  are controlled in a forced way irrespective of the inherent resonant frequency of the ultrasonic transducer  215  to switch the output voltage across the ultrasonic transducer  215 . The forced frequency mode therefore allows the H-bridge  334  to drive the ultrasonic transducer  215 , which has a resonant frequency f1 at different frequency f2. 
     Driving an ultrasonic transducer at a frequency which is different from its resonant frequency may be appropriate in order to adapt the operation to different applications. For example, it may be appropriate to drive an ultrasonic transducer at a frequency which is slightly off the resonance frequency (for mechanical reasons to prevent mechanical damage to the transducer). Alternatively, it may be appropriate to drive an ultrasonic transducer at a low frequency but the ultrasonic transducer has, because of its size, a different native resonance frequency. 
     The driver device  202  controls the bridge IC  301  to drive the ultrasonic transducer  215  in the forced frequency mode in response to the configuration of the driver device  202  for a particular application or a particular ultrasonic transducer. For instance, the driver device  202  may be configured to operate in the forced frequency mode when the resonant circuit device  201  is being used for a particular application, such as generating a mist from a liquid of a particular viscosity containing a drug for delivery to a user. 
     Native Frequency Mode (NFM) 
     The following native frequency mode of operation is a significant development and provides benefits in improved accuracy and efficiency over conventional ultrasonic drivers that are available on the IC market today. 
     The native frequency mode of operation follows the same switching sequence as described above but the timing of the zero output portion of the sequence is adjusted to minimise or avoid problems that can occur due to current spikes in the forced frequency mode operation. These current spikes occur when the voltage across the ultrasonic transducer  215  is switched to its opposite voltage polarity. An ultrasonic transducer which comprises a piezoelectric crystal has an electrical equivalent circuit which incorporates a parallel connected capacitor (e.g. see the piezo model in  FIG.  10   ). If the voltage across the ultrasonic transducer is hard-switched from a positive voltage to a negative voltage, due to the high dV/dt there can be a large current flow current flow as the energy stored in the capacitor dissipates. 
     The native frequency mode avoids hard switching the voltage across the ultrasonic transducer  215  from a positive voltage to a negative voltage (and vice versa). Instead, prior to applying the reversed voltage, the ultrasonic transducer  215  (piezoelectric crystal) is left free-floating with zero voltage applied across its terminals for a free-float period. The PMIC  300  sets the drive frequency of the bridge IC  301  such that the bridge  334  sets the free-float period such that current flow inside the ultrasonic transducer  215  (due to the energy stored within the piezoelectric crystal) reverses the voltage across the terminals of the ultrasonic transducer  215  during the free-float period. 
     Consequently, when the H-bridge  334  applies the negative voltage at the terminals of the ultrasonic transducer  215  the ultrasonic transducer  215  (the capacitor in the equivalent circuit) has already been reverse charged and no current spikes occur because there is no high dV/dt. 
     It is, however, to be appreciated that it takes time for the charge within the ultrasonic transducer  215  (piezoelectric crystal) to build up when the ultrasonic transducer  215  is first activated. Therefore, the ideal situation in which the energy within the ultrasonic transducer  215  is to reverse the voltage during the free-float period occurs only after the oscillation inside the ultrasonic transducer  215  has built up the charge. To accommodate for this, when the bridge IC  301  activates the ultrasonic transducer  215  for the first time, the PMIC  300  controls the power delivered through the H-bridge  334  to the ultrasonic transducer  215  to a first value which is a low value (e.g. 5 V), The PMIC  300  then controls the power delivered through the H-bridge  334  to the ultrasonic transducer  215  to increase over a period of time to a second value (e.g. 15 V) which is higher than the first value in order to build up the energy stored within the ultrasonic transducer  215 . Current spikes still occur during this ramp of the oscillation until the current inside the ultrasonic transducer  215  developed sufficiently. However, by using a low first voltage at start up those current spikes are kept sufficiently low to minimise the impact on the operation of the ultrasonic transducer  215 . 
     In order to implement the native frequency mode, the driver device  202  controls the frequency of the oscillator  315  and the duty cycle (ratio of turn-on time to free-float time) of the AC drive signal output from the H-bridge  334  with high precision. In this example, the driver device  202  performs three control loops to regulate the oscillator frequency and the duty cycle such that the voltage reversal at the terminals of the ultrasonic transducer  215  is as precise as possible and current spikes are minimised or avoided as far as possible. The precise control of the oscillator and the duty cycle using the control loops is a significant advance in the field of IC ultrasonic drivers. 
     During the native frequency mode of operation, the current sensor  335  senses the current flowing through the ultrasonic transducer  215  (resonant circuit) during the free-float period. The digital state machine  337  adapts the timing signals to switch on either the first switch T1 or the second switch T2 when the current sensor  335  senses that the current flowing through the ultrasonic transducer  215  (resonant circuit) during the free-float period is zero. 
       FIG.  12    of the accompanying drawings shows the oscillator voltage waveform  347  (V(osc)), a switching waveform  348  resulting from the turn-on and turn-off the left hand side high switch T1 of the H-bridge  334  and a switching waveform  349  resulting from the turn-on and turn-off the right hand side high switch T2 of the H-bridge  334 . For an intervening free-float period  350 , both high switches T1, T2 of the H-bridge  334  are turned off (free-floating phase). The duration of the free-float period  350  is controlled by the magnitude of the free-float control voltage  351  (Vphioff). 
       FIG.  13    of the accompanying drawings shows the voltage waveform  352  at a first terminal of the ultrasonic transducer  215  (the voltage waveform is reversed at the second terminal of the ultrasonic transducer  215 ) and the piezo current  353  flowing through the ultrasonic transducer  215 . The piezo current  353  represents an (almost) ideal sinusoidal waveform (this is never possible in the forced frequency mode or in any bridge in the IC market). 
     Before the sinusoidal wave of the piezo current  353  reaches zero, the left hand side high switch T1 of the H-bridge  334  is turned off (here, the switch T1 is turned off when the piezo current  353  is approximately 6 A). The remaining piezo current  353  which flows within the ultrasonic transducer  215  due to the energy stored in the ultrasonic transducer  215  (the capacitor of the piezo equivalent circuit) is responsible for the voltage reversal during the free-float period  350 . The piezo current  353  decays to zero during the free-float period  350  and into negative current flow domain thereafter. The terminal voltage at the ultrasonic transducer  215  drops from the supply voltage (in this case 19 V) to less than 2 V and the drop comes to a stop when the piezo current  353  reaches zero. This is the perfect time to turn on the low-side switch T3 of the H-bridge  334  in order to minimise or avoid a current spike. 
     Compared to the forced frequency mode described above, the native frequency mode has at least three advantages:
         1. The current spike associated with hard switching of the package capacitor is significantly reduced or avoided completely.   2. Power loss due to hard switching is almost eliminated.   3. Frequency is regulated by the control loops and will be kept close to the resonance of the piezo crystal (i.e. the native resonance frequency of the piezo crystal).       

     In the case of the frequency regulation by the control loops (advantage  3  above), the PMIC  300  starts by controlling the bridge IC  301  to drive the ultrasonic transducer  215  at a frequency above the resonance of the piezo crystal. The PMIC  300  then controls the bridge IC  301  to that the frequency of the AC drive signal decays/reduces during start up. As soon as the frequency approaches resonance frequency of the piezo crystal, the piezo current will develop/increase rapidly. Once the piezo current is high enough to cause the desired voltage reversal, the frequency decay/reduction is stopped by the PMIC  300 . The control loops of the PMIC  300  then take over the regulation of frequency and duty cycle of the AC drive signal. 
     In the forced frequency mode, the power delivered to the ultrasonic transducer  215  is controlled through the duty cycle and/or a frequency shift and/or by varying the supply voltage. However, in this example in the native frequency mode the power delivered to the ultrasonic transducer  215  controlled only through the supply voltage. 
     In this example, during a setup phase of operation of the driver device, the bridge IC  301  is configured to measure the length of time taken for the current flowing through the ultrasonic transducer  215  (resonant circuit) to fall to zero when the first switch T 1  and the second switch T 2  are turned off and the third switch T 3  and the fourth switch T 4  are turned on. The bridge IC  301  then sets the length of time of the free-float period to be equal to the measured length of time. 
     Referring now to  FIG.  14    of the accompanying drawings, the PMIC  300  and the bridge IC  301  of this example are designed to work together as a companion chip set. The PMIC  300  and the bridge IC  301  are connected together electrically for communication with one another. In this example, there are interconnections between the PMIC  300  and the bridge IC  301  which enable the following two categories of communication:
         1. control signals   2. feedback signals       

     The connections between the PHASE_A and PHASE_B pins of the PMIC  300  and the bridge IC  301  carry the PWM modulated control signals which drive the H-bridge  334 . The connection between the EN_BR pins of the PMIC  300  and the bridge IC  301  carries the EN_BR control signal which triggers the start of the H-bridge  334 . The timing between the PHASE_A, PHASE_B and EN_BR control signals is important and handled by the digital bridge control of the PMIC  300 . 
     The connections between the CS, OC and OT pins of the PMIC  300  and the bridge IC  301  carry CS (current sense), OC (over current) and OT (over temperature) feedback signals from the bridge IC  301  back to the PMIC  300 . Most notably, the CS (current sense) feedback signal comprises a voltage equivalent to the rms current flowing through the ultrasonic transducer  215  which is measured by the current sensor  335  of the bridge IC  301 . 
     The OC (over current) and OT (over temperature) feedback signals are digital signals indicating that either an over current or an over voltage event has been detected by the bridge IC  301 . In this example, the thresholds for the over current and over temperature are set with an external resistor. Alternatively, the thresholds can also be dynamically set in response to signals passed to the OC_REF pin of the bridge IC  301  from one of the two DAC channels VDAC0, VDAC1 from the PMIC  300 . 
     In this example, the design of the PMIC  300  and the bridge IC  301  allow the pins of these two integrated circuits to be connected directly to one another (e.g. via copper tracks on a PCB) so that there is minimal or no lag in the communication of signals between the PMIC  300  and the bridge IC  301 . This provides a significant speed advantage over conventional bridges in the IC market which are typically controlled by signals via a digital communications bus. For example, a standard I2C bus is clocked at only 400 kHz, which is too slow for communicating data sampled at the high clock speeds of up to 5 MHz of examples of this disclosure. 
     While examples of this disclosure have been described above in relation to the microchip hardware, it is to be appreciated that other examples of this disclosure comprise a method of operating the components and subsystems of each microchip to perform the functions described herein. For instance, the methods of operating the PMIC  300  and the bridge IC  301  in either the forced frequency mode or the native frequency mode. 
     Referring now to  FIG.  15    of the accompanying drawings, the optional OTP IC  242  comprises a power on reset circuit (POR)  354 , a bandgap reference (BG)  355 , a cap-less low dropout regulator (LDO)  356 , a communication (e.g. I2C) interface  357 , a one-time programmable memory bank (eFuse)  358 , an oscillator  359  and a general purpose input-output interface  360 . The OTP IC  242  also comprises a digital core  361  which includes a cryptographic authenticator. In this example, the cryptographic authenticator uses the Elliptic Curve Digital Signature Algorithm (ECDSA) for encrypting/decrypting data stored within the OTP IC  242  as well as data transmitted to and from the OTP IC  242 . 
     The POR  354  ensures that the OTP IC  242  starts up properly only if the supply voltage is within a predetermined range. If the supply voltage is outside the predetermined range, the POR  354  resets the OTP IC  242  and waits until the supply voltage is within the predetermined range. 
     The BG  355  provides precise reference voltages and currents to the LDO  356  and to the oscillator  359 . The LDO  356  supplies the digital core  361 , the communication interface  357  and the eFuse memory bank  358 . 
     The OTP IC  242  is configured to operate in at least the following modes:
         Fuse Programming (Fusing): During efuse programming (programming of the one time programmable memory) a high current is required to burn the relevant fuses within the eFuse memory bank  358 . In this mode higher bias currents are provided to maintain gain and bandwidth of the regulation loop.   Fuse Reading: In this mode a medium level current is required to maintain efuse reading within the eFuse memory bank  358 . This mode is executed during the startup of the OTP IC  242  to transfer the content of the fuses to shadow registers. In this mode the gain and bandwidth of the regulation loop is set to a lower value than in the Fusing Mode.   Normal Operation: In this mode the LDO  356  is driven in a very low bias current condition to operate the OTP IC  242  with low power so that the OTP IC  242  consumes as little power as possible.       

     The oscillator  359  provides the required clock for the digital core/engine  361  during testing (SCAN Test), during fusing and during normal operation. The oscillator  359  is trimmed to cope with the strict timing requirements during the fusing mode. 
     In this example, the communication interface  357  is compliant with the FM+ specification of the I2C standard but it also complies with &amp;ow and fast mode. The OTP IC  242  uses the communication interface  357  to communicate with the driver device  202  (the Host) for data and key exchange. 
     The digital core  361  implements the control and communication functionality of the OTP IC  242 . The cryptographic authenticator of the digital core  361  enables the OTP IC  242  to authenticate itself (e.g. using ECDSA encrypted messages) with the driver device  202  (e.g. for a particular application) to ensure that the OTP IC  242  is genuine and that the OTP IC  242  is authorised to connect to the driver device  202  (or another product). 
     With reference to  FIG.  16    of the accompanying drawings, the OTP IC  242  performs the following PKI procedure in order to authenticate the OTP IC  242  for use with a Host (e.g. the driver device  202 ):
         1. Verify Signer Public Key: The Host requests the Manufacturing Public key and Certificate. The Host verifies the certificate with the Authority Public key.   2. Verify Device Public Key: If the verification is successful, the Host requests the Device Public key and Certificate. The Host verifies the certificate with the Manufacturing Public key.   3. Challenge—Response: If the verification is successful, the Host creates a random number challenge and sends it to the Device. The End Product signs the random number challenge with the Device Private key.   4. The signature is sent back to the Host for verification using the Device Public key.       

     If all steps of the authentication procedure complete successfully then the Chain of Trust has been verified back to the Root of Trust and the OTP IC  242  is successfully authenticated for use with the Host. However, if any of the steps of the authentication procedure fail then the OTP IC  242  is not authenticated for use with the Host and use of the device incorporating the OTP IC  242  is restricted or prevented. 
     The foregoing outlines features of several examples or embodiments so that those of ordinary skill in the art may better understand various aspects of the present disclosure. Those of ordinary skill in the art should appreciate that they may readily use the present disclosure as a basis for designing or modifying other processes and structures for carrying out the same purposes and/or achieving the same advantages of various examples or embodiments introduced herein. Those of ordinary skill in the art should also realise that such equivalent constructions do not depart from the spirit and scope of the present disclosure, and that they may make various changes, substitutions, and alterations herein without departing from the spirit and scope of the present disclosure. 
     Although the subject matter has been described in language specific to structural features or methodological acts, it is to be understood that the subject matter of the appended claims is not necessarily limited to the specific features or acts described above. Rather, the specific features and acts described above are disclosed as example forms of implementing at least some of the claims. 
     Various operations of examples or embodiments are provided herein. The order in which some or all of the operations are described should not be construed to imply that these operations are necessarily order dependent. Alternative ordering will be appreciated having the benefit of this description. Further, it will be understood that not all operations are necessarily present in each embodiment provided herein. Also, it will be understood that not all operations are necessary in some examples or embodiments. 
     Moreover, “exemplary” is used herein to mean serving as an example, instance, illustration, etc., and not necessarily as advantageous. As used in this application, or is intended to mean an inclusive “or” rather than an exclusive or. In addition, “a” and “an” as used in this application and the appended claims are generally be construed to mean “one or more” unless specified otherwise or clear from context to be directed to a singular form. Also, at least one of A and B and/or the like generally means A or B or both A and B. Furthermore, to the extent that “includes”, “having”, “has”, “with”, or variants thereof are used, such terms are intended to be inclusive in a manner similar to the term “comprising”. Also, unless specified otherwise, “first,” “second,” or the like are not intended to imply a temporal aspect, a spatial aspect, an ordering, etc. Rather, such terms are merely used as identifiers, names, etc. for features, elements, items, etc. For example, a first element and a second element generally correspond to element A and element B or two different or two identical elements or the same element. 
     Also, although the disclosure has been shown and described with respect to one or more implementations, equivalent alterations and modifications will occur to others of ordinary skill in the art based upon a reading and understanding of this specification and the annexed drawings. The disclosure comprises all such modifications and alterations and is limited only by the scope of the following claims. In particular regard to the various functions performed by the above described features (e.g., elements, resources, etc.), the terms used to describe such features are intended to correspond, unless otherwise indicated, to any features which performs the specified function of the described features (e.g., that is functionally equivalent), even though not structurally equivalent to the disclosed structure. In addition, while a particular feature of the disclosure may have been disclosed with respect to only one of several implementations, such feature may be combined with one or more other features of the other implementations as may be desired and advantageous for any given or particular application. 
     Examples or embodiments of the subject matter and the functional operations described herein can be implemented in digital electronic circuitry, or in computer software, firmware, or hardware, including the structures disclosed in this specification and their structural equivalents, or in combinations of one or more of them. 
     Some examples or embodiments are implemented using one or more modules of computer program instructions encoded on a computer-readable medium for execution by, or to control the operation of, a data processing apparatus. The computer-readable medium can be a manufactured product, such as hard drive in a computer system or an embedded system. The computer-readable medium can be acquired separately and later encoded with the one or more modules of computer program instructions, such as by delivery of the one or more modules of computer program instructions over a wired or wireless network. The computer-readable medium can be a machine-readable storage device, a machine-readable storage substrate, a memory device, or a combination of one or more of them. 
     The terms “computing device” and “data processing apparatus” encompass all apparatus, devices, and machines for processing data, including by way of example a programmable processor, a computer, or multiple processors or computers. The apparatus can include, in addition to hardware, code that creates an execution environment for the computer program in question, e.g., code that constitutes processor firmware, a protocol stack, a database management system, an operating system, a runtime environment, or a combination of one or more of them. In addition, the apparatus can employ various different computing model infrastructures, such as web services, distributed computing and grid computing infrastructures. 
     The processes and logic flows described in this specification can be performed by one or more programmable processors executing one or more computer programs to perform functions by operating on input data and generating output. 
     Processors suitable for the execution of a computer program include, by way of example, both general and special purpose microprocessors, and any one or more processors of any kind of digital computer. Generally, a processor will receive instructions and data from a read-only memory or a random access memory or both. The essential elements of a computer are a processor for performing instructions and one or more memory devices for storing instructions and data. Generally, a computer will also include, or be operatively coupled to receive data from or transfer data to, or both, one or more mass storage devices for storing data, e.g., magnetic, magneto-optical disks, or optical disks. However, a computer need not have such devices. Devices suitable for storing computer program instructions and data include all forms of non-volatile memory, media and memory devices. 
     When used in this specification and claims, the terms “comprises” and “comprising” and variations thereof mean that the specified features, steps or integers are included. The terms are not to be interpreted to exclude the presence of other features, steps or components. 
     The invention may also broadly consist in the parts, elements, steps, examples and/or features referred to or indicated in the specification individually or collectively in any and all combinations of two or more said parts, elements, steps, examples and/or features. 
     In particular, one or more features in any of the embodiments described herein may be combined with one or more features from any other embodiment(s) described herein. 
     Protection may be sought for any features disclosed in any one or more published documents referenced herein in combination with the present disclosure. 
     REPRESENTATIVE FEATURES 
     Representative features are set out in the following clauses, which stand alone or may be combined, in any combination, with one or more features disclosed in the text and/or drawings of the specification. 
     1. A microchip for driving a resonant circuit, wherein the resonant circuit is an LC tank, an antenna or a piezoelectric transducer, and wherein the microchip is a single unit which comprises a plurality of interconnected embedded components and subsystems comprising:
         an oscillator which generates:
           a main clock signal,   a first phase clock signal which is high for a first time during the positive half-period of the main clock signal and low during the negative half-period of the main clock signal, and   a second phase clock signal which is high for a second time during the negative half-period of the main clock signal and low during the positive half-period of the main clock signal, wherein the phases of the first phase clock signal and the second phase clock signal are centre aligned;   
           a pulse width modulation (PWM) signal generator subsystem comprising:
           a delay locked loop which generates a double frequency clock signal using the first phase clock signal and the second phase clock signal, the double frequency clock signal being double the frequency of the main clock signal, wherein the delay locked loop controls the rising edge of the first phase clock signal and the second phase clock signal to be synchronous with the rising edge of the double frequency clock signal, and wherein the delay locked loop adjusts the frequency and the duty cycle of the first phase clock signal and the second phase clock signal in response to a driver control signal to produce a first phase output signal and a second phase output signal, wherein the first phase output signal and the second phase output signal are configured to drive an H-bridge circuit to generate an AC drive signal to drive the resonant circuit;   a first phase output signal terminal which outputs the first phase output signal to the H-bridge circuit;   a second phase output signal terminal which outputs the second phase output signal to the H-bridge circuit;   a feedback input terminal which receives a feedback signal from the H-bridge circuit;   
           an analogue to digital converter (ADC) subsystem comprising:
           a plurality of ADC input terminals which receive a plurality of respective analogue signals, wherein one ADC input terminal of the plurality of ADC input terminals is connected to the feedback input terminal such that the ADC subsystem receives the feedback signal from the H-bridge circuit, and wherein the ADC subsystem samples analogue signals received at the plurality of ADC input terminals at a sampling frequency which is proportional to the frequency of the main clock signal and the ADC subsystem generates ADC digital signals using the sampled analogue signals;   
           a digital processor subsystem which receives the ADC digital signals from the ADC subsystem and processes the ADC digital signals to generate the driver control signal, wherein the digital processor subsystem communicates the driver control signal to the PWM signal generator subsystem to control the PWM signal generator subsystem; and   a digital to analogue converter (DAC) subsystem comprising:
           a digital to analogue converter (DAC) which converts a digital control signal generated by the digital processor subsystem into an analogue voltage control signal to control a voltage regulator circuit which generates a voltage for modulation by the H-bridge circuit; and   a DAC output terminal which outputs the analogue voltage control signal to control the voltage regulator circuit to generate a predetermined voltage for modulation by the H-bridge circuit to drive the resonant circuit in response to feedback signals which are indicative of the operation of the resonant circuit.   
               

     2. The microchip of clause 1, wherein the oscillator generates the main clock signal at a frequency of 50 kHz to 105 MHz. 
     3. The microchip of clause 1, wherein the microchip further comprises:
         a frequency divider which is connected to the oscillator to receive the main clock signal from the oscillator, the frequency divider dividing the main clock signal by a predetermined divisor amount and outputting the frequency reference signal to the delay locked loop.       

     4. The microchip of clause 1, wherein the delay locked loop comprises a plurality of delay lines connected end to end, wherein the total delay of the delay lines is equal to the period of the main clock signal. 
     5. The microchip of clause 4, wherein the delay locked loop adjusts the duty cycle of the first phase clock signal and the second phase clock signal in response to the driver control signal by varying the delay of each delay line in the delay locked loop. 
     6. The microchip of clause 1, wherein the feedback input terminal receives a feedback signal which is indicative of a parameter of the operation of the H-bridge circuit or AC drive signal when the H-bridge circuit is driving the resonant circuit with the AC drive signal 
     7. The microchip of clause 1, wherein the feedback input terminal receives a feedback signal from the H-bridge circuit in the form of a voltage which indicative of an rms current of an AC drive signal which is driving the resonant circuit. 
     8. The microchip of clause 1, wherein the ADC subsystem comprises a plurality of further ADC input terminals which receive feedback signals which are indicative of at least one of the voltage of a battery connected to the microchip or the voltage of a battery charger connected to the microchip. 
     9. The microchip of clause 1, wherein the microchip further comprises:
         a temperature sensor which is embedded within the microchip, wherein the temperature sensor generates a temperature signal which is indicative of the temperature of the microchip, and wherein the temperature signal is received by a further ADC input terminal of the ADC subsystem and the temperature signal is sampled by the ADC.       

     10, The microchip of clause 1, wherein the ADC subsystem samples signals received at the plurality of ADC input terminals sequentially with each signal being sampled by the ADC subsystem a respective predetermined number of times. 
     11. The microchip of clause 1, wherein the microchip further comprises:
         a battery charging subsystem which controls the charging of an external battery which is connected to the microchip.       

     12. The microchip of clause 1, wherein the DAC subsystem comprises:
         a further digital to analogue converter (DAC) which converts a further digital control signal generated by the digital processor subsystem into a further analogue voltage control signal to control the voltage regulator circuit.       

     13. A microchip for driving a resonant circuit, wherein the resonant circuit is an LC tank, an antenna or a piezoelectric transducer, and wherein the microchip is a single unit which comprises a plurality of interconnected embedded components and subsystems comprising:
         a first power supply terminal;   a second power supply terminal;   an H-bridge circuit which incorporates a first switch, a second switch, a third switch and a fourth switch, wherein:
           the first switch and the third switch are connected in series between the first power supply terminal and the second power supply terminal;   a first output terminal is connected electrically between the first switch and the third switch,   the second switch and the fourth switch are connected in series between the first power supply terminal and the second power supply terminal, and   a second output terminal is connected electrically between the second switch and the fourth switch;   
           a first phase terminal which receives a first phase output signal from a pulse width modulation (PWM) signal generator;   a second phase terminal which receives a second phase output signal from the PWM signal generator;   a digital state machine which generates timing signals based on the first phase output signal and the second phase output signal and outputs the timing signals to the switches of the H-bridge circuit to control the switches to turn on and off in a sequence such that the H-bridge circuit outputs an AC drive signal for driving the resonant circuit, wherein the sequence comprises a free-float period in which the first switch and the second switch are turned off and the third switch and the fourth switch are turned on in order to dissipate energy stored by the resonant circuit;   a current sensor which incorporates:
           a first current sense resistor which is connected in series between the first switch and the first power supply terminal;   a first voltage sensor which measures the voltage drop across the first current sense resistor and provides a first voltage output which is indicative of the current flowing through the first current sense resistor;   a second current sense resistor which is connected in series between the second switch and the first power supply terminal;   a second voltage sensor which measures the voltage drop across the second current sensor resistor and provides a second voltage output which is indicative of the current flowing through the second current sense resistor; and   a current sensor output terminal which provides an rms output voltage relative to ground which is equivalent to the first voltage output and the second voltage output,   
           wherein the rms output voltage is indicative of an rms current flowing through the first switch or the second switch and the current flowing through the resonant circuit which is connected between the first output terminal and the second output terminal.       

     14. The microchip of clause 13, wherein the H-bridge circuit is configured to output a power of 22 W to 50 W to the resonant circuit which is connected to the first output terminal and the second output terminal. 
     15. The microchip of clause 13, wherein the microchip further comprises:
         a temperature sensor which is embedded within the microchip, wherein the temperatures sensor measures the temperature of the microchip and disables at least part of the microchip in the event that the temperature sensor senses that the microchip is at a temperature which is in excess of a predetermined threshold.       

     16. An apparatus for driving a resonant circuit, wherein the resonant circuit is an LC tank, an antenna or a piezoelectric transducer, the apparatus comprising:
         a first microchip, wherein the first microchip is a single unit which comprises a plurality of interconnected embedded components and subsystems comprising:   a first power supply terminal;   a second power supply terminal;   an H-bridge circuit which incorporates a first switch, a second switch, a third switch and a fourth switch, wherein:
           the first switch and the third switch are connected in series between the first power supply terminal and the second power supply terminal;   a first output terminal is connected electrically between the first switch and the third switch,   the second switch and the fourth switch are connected in series between the first power supply terminal and the second power supply terminal, and   a second output terminal is connected electrically between the second switch and the fourth switch;   
           a first phase terminal which receives a first phase output signal from a pulse width modulation (PWM) signal generator subsystem;   a second phase terminal which receives a second phase output signal from the PWM signal generator;   a digital state machine which generates timing signals based on the first phase output signal and the second phase output signal and outputs the timing signals to the switches of the H-bridge circuit to control the switches to turn on and off in a sequence such that the H-bridge circuit outputs an AC drive signal to the resonant circuit to drive the resonant circuit to generate and transmit the ultrasonic waves, wherein the sequence comprises a free-float period in which the first switch and the second switch are turned off and the third switch and the fourth switch are turned on in order to dissipate energy stored by the resonant circuit;   a current sensor which incorporates:
           a first current sense resistor which is connected in series between the first switch and the first power supply terminal;   a first voltage sensor which measures the voltage drop across the first current sense resistor and provides a first voltage output which is indicative of the current flowing through the first current sense resistor;   a second current sense resistor which is connected in series between the second switch and the first power supply terminal;   a second voltage sensor which measures the voltage drop across the second current sensor resistor and provides a second voltage output which is indicative of the current flowing through the second current sense resistor; and   a current sensor output terminal which provides an rms output voltage relative to ground which is equivalent to the first voltage output and the second voltage output,   
           wherein the rms output voltage is indicative of an rms current flowing through the first switch or the second switch and the current flowing through the resonant circuit which is connected between the first output terminal and the second output terminal; and   a second microchip connected to the first microchip to control the H-bridge circuit to generate the AC drive signal, wherein the second microchip is a single unit which comprises a plurality of interconnected embedded components and subsystems comprising:   an oscillator which generates:
           a main clock signal,   a first phase clock signal which is high for a first time during the positive half-period of the main clock signal and low during the negative half-period of the main clock signal, and   a second phase clock signal which is high for a second time during the negative half-period of the main clock signal and low during the positive half-period of the main clock signal, wherein the phases of the first phase dock signal and the second phase dock signal are centre aligned;   
           the pulse width modulation PWM signal generator subsystem, wherein the PWM signal generator subsystem comprises:
           a delay locked loop which generates a double frequency clock signal using the first phase clock signal and the second phase clock signal, the double frequency clock signal being double the frequency of the main clock signal, wherein the delay locked loop controls the rising edge of the first phase clock signal and the second phase clock signal to be synchronous with the rising edge of the double frequency clock signal, and wherein the delay locked loop adjusts the frequency and the duty cycle of the first phase clock signal and the second phase clock signal in response to a driver control signal to produce a first phase output signal and a second phase output signal, wherein the first phase output signal and the second phase output signal are configured to drive the H-bridge circuit to generate an AC drive signal to drive the resonant circuit;   a first phase output signal terminal which outputs the first phase output signal to the H-bridge circuit;   a second phase output signal terminal which outputs the second phase output signal to the H-bridge circuit;   a feedback input terminal which receives a feedback signal from the H-bridge circuit, the feedback signal being indicative of a parameter of the operation of the H-bridge circuit or the AC drive signal when the H-bridge circuit is driving the resonant circuit with the AC drive signal;   
           an analogue to digital converter (ADC) subsystem comprising:
           a plurality of ADC input terminals which receive a plurality of respective analogue signals, wherein one ADC input terminal of the plurality of ADC input terminals is connected to the feedback input terminal such that the ADC subsystem receives the feedback signal from the H-bridge circuit, and wherein the ADC subsystem samples analogue signals received at the plurality of ADC input terminals at a sampling frequency which is proportional to the frequency of the main clock signal and the ADC subsystem generates ADC digital signals using the sampled analogue signals;   
           a digital processor subsystem which receives the ADC digital signals from the ADC subsystem and processes the ADC digital signals to generate the driver control signal, wherein the digital processor subsystem communicates the driver control signal to the PWM signal generator subsystem to control the PWM signal generator subsystem; and   a digital to analogue converter (DAC) subsystem comprising:
           a digital to analogue converter (DAC) which converts a digital control signal generated by the digital processor subsystem into an analogue voltage control signal to control a voltage regulator circuit which generates a voltage for modulation by the H-bridge circuit; and   
               

     a DAC output terminal which outputs the analogue voltage control signal to control the voltage regulator circuit to generate a predetermined voltage for modulation by the H-bridge circuit to drive the resonant circuit in response to feedback signals which are indicative of the operation of the resonant circuit. 
     17. The apparatus of clause 16, wherein the apparatus further comprises:
         a boost converter circuit which is configured to increase the voltage of a power supply to a boost voltage in response to the analogue voltage output signal from the DAC output terminal, wherein the boost converter circuit provides the boost voltage at the first power supply terminal such that the boost voltage is modulated by the switching of the switches of the H-bridge circuit.       

     18. The apparatus of clause 16, wherein the current sensor senses the current flowing through the resonant circuit during the free-float period and the digital state machine adapts the timing signals to switch on either the first switch or the second switch when the current sensor senses that the current flowing through the resonant circuit during the free-float period is zero. 
     19. The apparatus of clause 16, wherein, during a setup phase of operation of the apparatus, the second microchip:
         measures the length of time taken for the current flowing through the resonant circuit to fall to zero when the first switch and the second switch are turned off and the third switch and the fourth switch are turned on; and   sets the length of time of the free-float period to be equal to the measured length of time.       

     Although certain example embodiments of the invention have been described, the scope of the appended claims is not intended to be limited solely to these embodiments. The claims are to be construed literally, purposively, and/or to encompass equivalents.